Automatic frequency control method and apparatus therefor

Abstract: An automatic frequency control apparatus includes a sampling portion (30) for sampling the signal transmitted according to a phase-shift keying method, a phase difference detector (32) for detecting the phase difference between the currently sampled signal (R k ) and the immediately preceding sampled signal (R k-1 ) , and a determinator (34, 36) for determining a reference phase which is the closest to the phase of the transmitted signal using a quantization characteristic of phase during transmission. The frequency offset generated by the distance between the frequency of a carrier wave and the local oscillation frequency of a receiver, or by the Doppler shift, in a MPSK communication method is determined as the phase difference between the detected phase difference information and the determined reference phase information. Since the frequency offset is detected using the signal sampled at the symbol rate, the cost of hardware constituting the apparatus can be reduced. The apparatus can be used for the automatic frequency control of a modem using all types of MPSK modulation.

Description: Background of the Invention The present invention relates to an automatic frequency control (AFC) method and apparatus therefor, and more particularly, though not exclusively, to an automatic frequency control method and apparatus therefor which are used for detecting a synchronous signal from a signal which is M-ary phase-shift keying modulated and transmitted. Digital transmission technology is widely used since it guarantees high quality transmission and improves frequency efficiency. With this technology, a receiver of a synchronous detecting method performs a signal demodulation process after detecting the synchronous signal from the signal which is transmitted after being modulated. In order to detect a synchronous signal from the received signal, the oscillation frequency of the local carrier wave in the receiver has to match the frequency of the modulated carrier wave and no phase error should be present. As a result, if this is achieved the original signal can be perfectly detected. To achieve this, a local carrier wave is synchronized with the carrier of the modulated signal. The conventional technology for detecting the synchronous signal from the received signal will be described, referring to Figures. 1, 2A and 2B. Figure 1 is a block diagram showing an automatic frequency control apparatus for conventional quadrature phase-shift keying (QPSK) demodulation. In Figure 1, the received signal S(t) is input to multipliers 11 and 12. Multipliers 11 and 12 multiply the received signals by first and second local carrier waves which are output from voltage-controlled oscillator (VCO) 19 and have a phase difference of 90°. Here, the first and second local carrier waves can be represented as 2 cos ω 1 t and 2 sin ω 1 t, respectively. The output signals of multipliers 11 and 12 are low-pass-filtered by low-pass filters (LPF) 13 and 14, respectively, to become baseband signals. When Gaussian noise is not included in the received signal, the received signal S(t) can be expressed as A cos [ω 0 t+(t)]. Low-pass filter 13 outputs an I-channel (in-phase) signal I(t) expressed as A cos [Δωt+(t)], and low-pass filter 14 outputs a Q-channel (quadrature) signal Q(t) expressed as A sin [Δωt+(t)]. Here, Δω=ω 1 -ω 0 . The output signals of low-pass filters 13 and 14 are digitized in analog-to-digital (A/D) converters 15 and 16, respectively, and are output as demodulated signals. At the same time, the digitized signals (I k and Q k ) are input to a frequency detector 17. The signals output from A/D converters 15 and 16 each have a phase (t) varied for every symbol period T b . For example, in quadrature phase-shift keying, phase (t) is 45°, 135°, -45° or -135° and is varied according to the bit stream information of a transmission side and symbol period T b . A frequency detector 17 receives the two channel signals I k and Q k , to generate frequency offset information V(t). The frequency offset is generated since the actual oscillation frequency (local carrier wave) of a receiver and the frequency of the received carrier wave do not coincide. Frequency offset information V(k) is input to VCO 19 through a loop filter 18. VCO 19 generates the first and second local carrier waves according to the input frequency offset information. The first and second local carrier waves have the same frequency and a phase difference of 90°, for generating the I- and Q-channel signals. The technology for detecting the above-described frequency offset information is discussed in "AFC Tracking Algorithms," IEEE Transactions on Communications, Vol. COM-32, No. 8, pp.935-947. Figure 2A shows a frequency detector using the cross product disclosed in the above reference. The I-channel signal I k , which is sampled and digitized, is applied to a delay circuit (D) 21 and a multiplier 24, and the Q-channel signal Q k , which is also sampled and digitized, is applied to a delay circuit 22 and a multiplier 23. Multiplier 23 multiplies the delayed I-channel signal by the input Q-channel signal, and multiplier 24 multiplies the delayed Q-channel signal by the input I-channel signal. A subtractor 25 subtracts the output signal of multiplier 24 from the output signal of multiplier 23 and generates frequency offset information V(k) which is determined according to a sampling period T s and symbol period T b . That is, when sampling period T s is the same as symbol period T b , i.e., T s =T b , subtractor 25 generates frequency offset information V(k) in accordance with the following. I k = A cos (ΔωkT s +  k ) I k-1 = A cos [Δω(k-1)T s +  k-1 ] Q k = A sin (ΔωkT s +  k ) Q k-1 = A sin [Δω(k-1)T s +  k-1 ] Q k I k -1 = A 2 2 sin(ΔωT s +  k -  k -1 ) + A 2 2 sin[Δω(2 k - 1) T s +  k +  k -1 ] I k Q k -1 = - A 2 2 sin(ΔωT s +  k -  k -1 ) + A 2 2 sin[Δω(2 k - 1)T s +  k +  k -1 ] ∴ V(k) = Q k I k-1 - I k Q k-1 = A 2 sin (ΔωT s +  k ) Here, it is assumed that Δω=ω 1 -ω 0 , (t)= k , kT s ≤t<(k+1)T s and  k = k - k-1 . On the other hand, when T s <T b (over-sampling: T b =nT s , where n>1), frequency offset information V(k) generated from subtractor 25 is expressed as the following formula (20). Here, when k is not equal to nl, then I k = A cos (ΔωkT s +  l ) I k-1 = A cos [Δω(k-1)T s +  l ] Q k = A sin (ΔωkT s +  l ) Q k-1 = A sin [Δω(k-1)T s +  l ] Q I k -1 = A 2 2 sin(ΔωT s ) + A 2 2 sin[Δω (2 k - 1)T s + 2 ℓ ] I k Q k -1 = - A 2 2 sin(ΔωT s ) + A 2 2 sin[Δω(2 k - 1)T + 2 ℓ ] ∴ V(k) = Q k I k-1 - I k Q k-1 = A 2 sin (ΔωT s ) and when k is equal to nl, then I k = A cos (ΔωkT s +  l ) I k-1 = A cos [Δω(k-1)T s +  l-1 ] Q k = A sin (ΔωkT s +  l ) Q k-1 = A sin [Δω(k-1)T s +  l-1 ] Q k I k -1 = A 2 2 sin(ΔωT s +  ℓ -  ℓ-1 ) + A 2 2 sin[Δω(2 k - 1)T s +  ℓ +  ℓ-1 ] I k Q k -1 = - A 2 2 sin(ΔωT s +  ℓ -  ℓ-1 ) + A 2 2 sin[Δω(2 k - 1)T s +  ℓ +  ℓ-1 ] ∴ V(k) = Q k I k-1 - I k Q k-1 = A 2 sin (ΔωT s + ℓ ) Therefore, V ( k ) = A 2 sin(ΔωT s ) for k ≠ n ℓ V ( k ) = A 2 sin(ΔωT s +  ℓ ) for k ≠ n ℓ wherein l=k/n and l is an integer not over [x]=x. Figure 2B shows a frequency detector using an arc tangent operation. An arc tangent portion 27 receives the signals (Q k and I k ) of two channels and performs the arc tangent operation by taking the Q-channel signal Q k as a numerator and the I-channel signal Ik as a denominator. The output signal of arc tangent portion 27 is input to a differentiator 28 which generates frequency offset information V(k) expressed by formulas 21 and 22. First, from formulas 1 and 3, F(k) is defined as follows when T s =T b : F(k) = tan -1 (Q k /I k ) = ΔωkT s +  k ∴ V(k) = F(k)-F(k-1) = Δω(k-k+1)T s + ( k - k-1 ) = ΔωT s +  k In the case of over-sampling and k≠nl, from formulas 8 and 10, we get F(k) = tan -1 (Q k /I k ) = ΔωkT s +  l and from formulas 9 and 11, we get F(k-1) = tan -1 (Q k-1 /I k-1 ) = Δω(k-1)T s +  l ∴ V(k) = F(k)-F(k-1) = ΔωT s On the other hand, in over-sampling where k=nl, from formulas 14 and 16, we get F(k) = tan -1 (Q k /I k ) = ΔωkT s +  l and from formulas 15 and 17, we get F(k-1) = tan -1 (Q k-1 /I k-1 ) = Δω(k-1)T s +  l-1 ∴ V(k) = F(k) - F(k-1) = ΔωT s +  l -  l-1 = ΔωT s +  l Therefore, V ( k ) = ΔωT s for k≠ n ℓ V ( k ) = ΔωT s +  ℓ for k≠ n ℓ wherein l=k/n and l is an integer not over [x]=x. The above formula 21 represents the output signal of differentiator 28 in the case where T s is equal to T b , and formula 22 represents the output signal of differentiator 28 in the case of over-sampling, that is, T s <T b and T b =nT s . Terms  k and  l , whose values are varied according to the transmitted information, are excluded from the formula, in order to detect the exact frequency. Here,  k and  l represent the difference between the phase information transmitted from the current symbol period and the phase information transmitted from immediately preceding symbol period, that is, the phase difference between symbols during transmission. Hereinafter,  k is referred to as transmission phase information. A QPSK demodulator having a phase difference detector for performing an arc tangent operation as above and performing an AFC function is disclosed in European Patent Application No. 0 526 836 A2. However, as shown in formulas 7 and 21, transmission phase information  k is included in the formula in the case where the sampling frequency is equal to the symbol rate. Thus, it is impossible to detect the exact frequency directly proportional to frequency offset Δω. Also, in the case of over-sampling, as shown in formulas 20 and 22, it is impossible to detect the exact frequency, since transmission phase information  l remains in a sample which undergoes symbol transition. Although the performance of frequency detection can be improved according to the degree of over-sampling, the cost of hardware for over-sampling is increased for symbol rates above 20MHz, such as in the case of a direct broadcasting satellite. This problem can also be generated in a differentiator AFC circuit or a discrete Fourier transform AFC circuit. EP-A-413 153 discloses a device for an angular modulated signal is quadrature demoldulated into first and second signal vectors by a local carrier signal having a frequency signal difference from a received carrier signal which is angular modulated by a modulating signal comprising symbols representative of binary one and zero at a symbol interval. In the modulating signal, a particular symbol sequence comprises the symbols in a predetermined manner and results in the first signal vector. A data symbol sequence comprises the symbols to represent data and results in the second signal vector. The first signal vector has inphase and quadrature phase amplitudes which are converted to phase values substantially at the symbol interval. An individual error is provided by comparing a phase variation between two phase values with a reference variation determined by the predetermined manner. Such individual errors are averaged into a phase error which is used in correcting the second signal vector. It is possible to compute the phase error by calculating the phase variation using a reference variation determined by the predetermined manner and the received carrier signal. Summary of the Invention It is an aim of embodiments of the present invention to provide an automatic frequency control method for automatically controlling frequency by obtaining a reference phase which is the closest to the phase of a transmitted signal using the quantization characteristic of the phase during transmission and the phase difference between the currently sampled signal and the immediately preceding sampled signal, after sampling the signal transmitted according to a phase-shift keying method, and detecting a frequency offset. It is another aim of embodiments of the present invention to provide an automatic frequency control apparatus suitable for performing the above method. According to the present invention there is provided an automatic frequency control method in which a MPSK-modulated signal is received, a frequency offset signal is generated using the phase information of the modulated signal, and the local oscillation frequency of a receiver is controlled on the basis of the frequency offset signal, comprises the steps of: sampling the received signal at a predetermined sampling frequency; generating a phase detection signal having the phase difference information between the currently sampled signal and the immediately preceding sampled signal; determining a reference phase which is the closest to the phase difference information of the phase detection signal among the reference phase values of the M-ary phase-shift keying of the modulated signal, as transmission phase information of the received signal; and generating a frequency offset signal corresponding to the phase difference between the phase difference information and the determined transmission phase information. Suitably, said phase detection signal generating step comprises a normalization step where a first phase detection signal U k is generated according to U k = R k R * k -1 R k x R k -1 = e j ( k +ΔωT) ≡ C k + jS k by using a currently sampled signal R k and an immediately preceding sampled signal R k-1 where  k represents the difference between phase information transmitted from the current symbol period and the phase information transmitted from the immediately preceding symbol period, and ΔωT is a frequency offset signal. Suitably, said phase detection signal generating step comprises a complex conjugation operation step where a second phase detection signal U' k is generated according to U' k = R k R* k-1 by using a currently sampled signal R k and an immediately preceding sampled signal R k-1 . Suitably, said frequency offset signal generating step comprises a first frequency offset signal generation step where a first frequency offset signal sin (ΔωT) is generated according to sin (ΔωT) = Im[U k e -jψi ] using said reference phase ψ i and said phase detection signal U k . Suitably, said frequency offset signal generating step further comprises a second frequency offset signal generation step where a second frequency offset signal ΔωT is generated using the phase of said first frequency offset signal sin (ΔωT). Suitably, in said transmission phase information determination step, when said phase difference information ∠U k of phase detection signal U k includes a second frequency offset signal ΔωT between -π/M and π/M determined according to the number M of the reference phase of the M-ary phase-shift keying, and when said phase difference information ∠U k , second frequency offset signal ΔωT and transmission phase information  k satisfy ΔωT = ∠U k -  k , a reference phase ψ i is determined as the reference phase which is the closest to said phase difference information. According to another aspect of the present invention there is provided an automatic frequency control apparatus in which a MPSK-modulated signal is received, a frequency offset signal is generated using the phase information of the modulated signal, and the local oscillation frequency of a receiver is controlled on the basis of the frequency offset signal, comprises: sampling means for sampling the received signal at a predetermined sampling frequency and outputting the sampled signal; phase difference detection means for receiving the currently sampled signal from the sampling means and for generating a phase detection signal having phase difference information between the currently sampled signal and the immediately preceding sampled signal stored therein; and means for receiving the phase detection signal to determine a reference phase which is the closest to the phase difference information of the phase detection signal among the reference phase values of the M-ary phase-shift keying of the modulated signal, as transmission phase information of the received signal, and for generating a frequency offset signal determined according to the phase difference between the phase difference information and the transmission phase information. Suitably, said phase difference detection means (32) comprises normalization means for receiving the currently sampled signal (R k ) and the immediately preceding sampled signal (R k-1 ) and generating said phase detection signal (U k ) . Suitably, said phase difference detection means comprises a complex conjugation operator for receiving the currently sampled signal (R k ) and the immediately preceding sampled signal (R k-1 ) and generating said phase detection signal (U k ). Suitably, said frequency offset signal generation means comprises: determination means for storing a plurality of critical values determined by said reference phase values, receiving said phase detection signal and outputting the reference phase value determined by said phase difference information of said phase detection signal and said plurality of critical values; and first frequency offset signal generator for receiving said transmission phase information and said phase detection signal and generating first frequency offset signal which is an imaginary number component generated through the complex conjugation operation between the received transmission phase information and phase detection signal. Suitably, said first frequency offset generator receives said reference phase signal e -jψi and said phase detection signal from said determination means and generates a first frequency offset signal sin (ΔωT) determined according to sin (ΔωT) = Im[U k e -jψi ] = S k cos ψ i - C k sin ψ i . Suitably, said automatic frequency control apparatus further comprises a second frequency offset signal generator for receiving said first frequency offset signal sin (ΔωT) and generating a second frequency offset signal ΔωT through an arc sine operation. Suitably, in said determination means, when said phase difference information ∠U k of said phase detection signal includes a second frequency offset signal ΔωT between -π/M and π/M determined according to the number M of reference phase of phase-shift keying, and when said phase difference information ∠U k , second frequency offset signal ΔωT and transmission phase information  k satisfy ΔωT = ∠U k -  k , a reference phase ψ i is determined as the reference phase which is the closest to said phase difference information. Suitably, said sampling means samples said received signal at the sampling frequency equal to the symbol rate. Suitably, said frequency offset signal generator comprises: a first sign generator for receiving the QPSK-modulated phase detection signal and generating a first sign signal which is the value of a sign function with respect to the difference obtained by subtracting a second component signal constituting said phase detection signal from first component signal constituting said phase detection signal; a second sign generator for generating a second sign signal which is the value of a sign function with respect to the sum of said first and second component signals; a subtractor for subtracting said second sign signal from said first sign signal; a first adder for summing said first and second sign signals; a third sign generator for outputting a third sign signal which is the sign function value of the signal output from said subtractor; a fourth sign generator for outputting a fourth sign signal which is the sign function value of the signal output from said adder; a first multiplier for multiplying said first component signal by the output signal of said third sign generator; a second multiplier for multiplying said second component signal by the output signal of said fourth sign generator; and a second adder for summing the output signals of said first and second multipliers and generating said first frequency offset signal. Suitably, said frequency offset signal generator comprises: a control signal generator for receiving the QPSK-modulated phase detection signal and generating a selection control signal determined according to a first sign function value with respect to the difference obtained by subtracting the second component signal constituting said phase detection signal from the first component signal constituting said phase detection signal and a second sign function value with respect to the sum of said first and second component signals, said control signal generator generating said selection control signal so that the first component signal is selected as said first frequency offset signal if the first and second sign function values are greater than zero, the inverted signal of the first component signal is selected as said first frequency offset signal if the first sign function value is less than zero and the second sign function value is greater than zero, the inverted signal of the second component signal is selected as said first frequency offset signal if the first and second sign function values are less than zero, and the second component signal is selected as said first frequency offset signal if the first sign function value is greater than zero and the second sign function value is less than zero; a first signal inverter for inverting and outputting said first component signal; a second signal inverter for inverting and outputting said second component signal; and selection means comprising four signal inputs connected for receiving said first and second component signals and the output signals of said first and second signal inverters, for selecting one of said four input signals according to said selection control signal and outputting said selected signal as said first frequency offset signal. Suitably, said automatic frequency control apparatus further comprises a second frequency offset signal generator for receiving said first frequency offset signal and generating a second frequency offset signal through an arc sign operation. Suitably, said first component signal is a real number component of said phase detection signal and said second component signal is an imaginary number component of said phase detection signal. Suitably, said sampling means samples said received signal at the sampling frequency equal to the symbol rate. Brief Description of the Drawings The above advantages of the present invention will become more apparent by describing in detail preferred embodiments thereof, by way of example only, with reference to the attached drawings in which: Figure 1 is a block diagram showing an automatic frequency control apparatus for conventional QPSK demodulation; Figures 2A and 2B are block diagrams of a conventional frequency detector; Figure 3 is a diagram showing the relation between transmission phase information and a reference phase signal according to the present invention; Figure 4 is a block diagram showing a frequency detector according to a preferred embodiment of the present invention; Figure 5 is a discriminator curve showing the frequency offset; Figures 6A and 6B are diagrams for illustrating how the transmission phase information is determined by the real and imaginary number components of the phase detection signal in the case of QPSK; Figure 7 is a block diagram showing the frequency detector according to another preferred embodiment of the present invention in the case of QPSK; and Figure 8 is a block diagram showing the frequency detector according to still another preferred embodiment of the present invention in the case of QPSK. Detailed Description of Preferred Embodiments First, referring to Figure 3, an automatic frequency control method will be described below. To sum up the automatic frequency control method according to the present invention, the transmitted signal is sampled at a symbol rate. If the sampling period is equal to the symbol period, the exact frequency is detected by eliminating the k term through the assumption of the transmission phase information. That is, transmission phase information  k is assumed by determining a reference phase ψ i among a plurality of reference phases for transmitting information of the M-ary phase-shift keying as a reference phase. Here, the signal modulation using phase-shift keying varies transmission phase information k, in order to transmit information. Therefore, according to the present invention, the frequency offset is detected so as to exactly perform the automatic frequency control by subtracting the assumed transmission phase information (reference phase) from the phase difference information between the currently sampled signal and the immediately preceding sampled signal. The automatic frequency control method will be described in detail below. If a signal modulated by M-ary phase-shift keying is transmitted, the receiver demodulates the received signal and the automatic frequency control apparatus samples the input signal. The size r k ' of a sampled signal R k is varied by noise. Since the sampled signal includes a frequency offset signal ΔωT, the sampled signal R k can be expressed as the following formula 23. R k = I k + jQ k = r ' k e j ( k +ΔωT k ) where T=T s =T b . As shown in the following formula 24, if normalization is performed using the currently sampled signal R k and the immediately preceding sampled signal R k-1 , the obtained signal has the phase difference information contained therein. Thus, the signal is called a phase detection signal U k which is obtained by compensating the magnitude of the signal distorted by noise. U k ≡ R k R * k -1 R k x R k -1 = r ' k e j ( k +ΔωT k ) r ' k -1 e - j [ ( k -1) +ΔωT ( k -1) ] r ' k r ' k -1 = e j [ k - ( k -1) +ΔωT] = e j ( k +ΔωT) ≡ C k + jS k As shown in formula 24, phase difference signal ∠U of phase detection signal U k always satisfies the following formulas 25 and 26. ∠U k =  k + ΔωT ΔωT = ∠U k -  k Sampled signal R k is the received signal, and phase detection signal U k can be calculated by the above formula 24. Therefore, if transmission phase information  k , which is varied according to the transmitted information, is known, frequency offset signal ΔωT can be calculated exactly. However, transmission phase information  k is varied by transmission information and can itself be used as transmission information in the case of differential encoding. Thus, if the frequency control apparatus does not operate smoothly, the receiver cannot exactly determine the transmission phase information. To solve the problem, the frequency offset can be detected by transmitting the reference phase information in a constant symbol interval. However, this method is inefficient in terms of signal transmission. Therefore, according to the present invention, the following method is suggested. Transmission phase information  k which is not a continuous value, is quantized by the reference phase of M-ary phase-shift keying, and is transmitted. During the transmission, transmission phase information  k is quantized to have reference phase values as shown in formula 27 below.  k ∈{ψ i |ψ i = 0,± 2 M π,± 4 M π,± 6 M π,...,± M - 2 M π,π} Therefore, the phase difference between the symbols can be detected from the sampled signal and the reference phase signal which is the closest to each of the above transmission phase information values can be detected even when a frequency offset exists. Figure 3 schematically shows the relation between the transmission phase information and the reference phase signal according to the present invention. In the case where the range for selecting the reference phase signal which is the closest to transmission phase information  k is determined according to the following formula 28, it is determined that reference phase signal e jψi and transmission phase information k within the range have the same information. -π/M < ΔωT < π/M Therefore, the automatic frequency control apparatus according to the present invention determines reference phase signal (e jψi of Figure 3) which is located between the boundary lines (dotted lines) of Figure 3 including the detected phase detection signal U k , as transmission phase information  k . A parameter M of the above formula 28 has the same value as the number of reference phase signal e jψi for phase-shift keying. If reference phase signal e jψi regarded as transmission phase information  k is determined, frequency offset signal ΔωT having the relation shown in the above formula 25 or 26 is calculated by the following formula 30 or 31. Frequency offset signal ΔωT is obtained using formula 29 for obtaining complex numbers A and B and the angle of two complex numbers. sin (ΔωT) = Im[U k e -jψi ] = Im[(C k +jS k )e -jψi ] = Im[(C k +jS k ) × ( cos ψ i - sin ψ i )] = S k cos ψ i - C k sin ψ i _ ΔωT ΔωT = sin -1 (S k cos ψ i -C k sin ψ i ) By the above formula 30, the frequency offset signal can be more simply detected compared with the case of using formula 31. The apparatus realized using the above formula 23 can detect the frequency offset signal having a negligible error. Next, the apparatus adopting the automatic frequency control method according to the present invention will be described in detail, referring to Figures 4-6. Figure 4 is a block diagram showing a frequency detector according to a preferred embodiment of the present invention. The automatic frequency control apparatus shown in Figure 4 comprises a sampling portion 30, a phase difference detector 32, a determiner 34, a first frequency offset signal generator 36, and a second frequency offset signal generator 38. Here, phase difference detector 32 may comprise normalization means employing the above formula 24 or, in cases where a large amount of calculation is needed, a complex conjugation operator employing the formula U k ' = R k R* k-1 . Now, the operation of the apparatus shown in Figure 4 will be described. Sampling portion 30 samples and outputs the signal which is received and demodulated. According to the present invention, a sampling frequency f s of the signal is set as the same as symbol rate T b so as to provide for less-expensive hardware. When sampled signal R k is input, phase difference detector 32 generates phase detection signal U k or U' k using the currently sampled signal R k and the immediately preceding sampled signal R k-1 being stored therein. Normalization means or complex conjugation operator selected by an external control signal (not shown) generates the phase detection signal at phase difference detector 32. The phase detection signal generated from phase difference detector 32 is applied to determiner 34 and first frequency offset signal generator 36. Since the same operation is performed in response to two phase detection signals U k and U' k , hereinafter, signal processing in response to only phase detection signal U k will be described. Determiner 34 determines transmission phase information k using the critical values stored therein. That is, the reference phase which is the closest to phase difference information ∠U k included in phase detection signal U k , among the reference phase values which the received signal may have, is determined as transmission phase information  k of the modulated signal. Thus, determiner 34 outputs reference phase signal e jψi having the transmission phase information. First frequency offset signal generator 36 receives phase detection signal U k and reference phase signal e jψi and generates first frequency offset signal sin (ΔωT). First frequency offset signal generator 36 is realized by the above formula (30) and generates the imaginary number component of the complex value obtained by complex-conjugation-operating the input reference phase signal e jψi and phase detection signal U k . When the actual circuit is realized, first frequency offset signal generator 36 is operated according to the expression S k cos ψ i -C k sin ψ i by receiving real and imaginary number components C k and S k of phase detection signal U k and reference phase information ψ i of determiner 34. First frequency offset signal sin(ΔωT) output from first frequency offset generator 36 is applied to second frequency offset generator 38. Second frequency offset generator 38 outputs the arc sine value of the input first frequency offset signal sin (ΔωT). That is, a second frequency offset signal ΔωT is generated. All outputs of first and second frequency offset signal generators 36 and 38 can be used for generating a local oscillation signal by using frequency offset information. Figure 5 graphically shows frequency offset information generated by the frequency offset signal generators. The characteristic curve of Figure 5 is obtained by a discriminator under the condition where |ΔωT| is less than π/M, |Δf| is less than f s /2M, and f s is equal to 1/T b . Curves A and B of Figure 5, obtained by the discriminator, are formed by first frequency offset signal generator 36 and second frequency offset signal generator 38, respectively. First frequency offset signal generator 36 generates non-linear discriminator curve A near boundary π/M. If non-linear discriminator curve A is a flat, increasing function, it is unnecessary to be linear such as curve B. Thus, the output signal of first frequency offset signal generator 36 can be used as frequency offset information in the actual apparatus. Here, if second frequency offset signal generator 38 is used, the frequency offset information can be detected with great accuracy. However, there is a problem of an increase in the calculation amount. Figures 6-8 show the case where the automatic frequency control method of the present invention is applied to QPSK modulation. In QPSK modulation, the value of parameter M is four and reference phases ψ 0 , ψ 1 , ψ 2 and ψ 3 are 0, π/2, π and -π/2, respectively, according to formula (27). The sum or difference between the real and imaginary number components C k and S k of phase detection signal U k has the relation shown in Figures 6A and 6B. Figure 6A represents sgn (C k -S k ) and Figure 6B represents sgn (C k +S k ). The sum or difference between the real and imaginary number components C k and S k is determined as one of the regions divided by the coordinate axis and dotted diagonal lines, according to the sign function. The sign function is defined as follows. sgn (x) = 1 for x>0 sgn (x) = 0 for x=0 sgn (x) = -1 for x<0 Reference phase information ψ i simultaneously corresponding to the regions determined in Figures 6A and 6B is transmission phase information  k . These relations are summarized as follows: (a) if C k -S k >0 and C k +S k >0, then ψ i =0; (b) if C k -S k <0 and C k +S k >0, then ψ i =π/2; (c) if C k -S k <0 and C k +S k <0, then ψ i =π; and (d) if C k -S k >0 and C k +S k <0 then, ψ i =-π/2. Frequency offset signal ΔωT is determined by the reference phase determined by the above relations and formula (30). That is, if ψ i is equal to zero, sin (ΔωT) is equal to S k ; if ψ i is equal to π /2, sin (ΔωT) is equal to -C k ; if ψ i is equal to π, sin (ΔωT) is equal to -S k ; and if ψ i is equal to -π/2, sin (ΔωT) is equal to C k . Summarizing the above, the following formula (32) is formed. sin(ΔωT) = C k sgn[sgn( C k - S k )sgn( C k + S k )] + S k sgn[sgn( C k - S k )] Figure 7 is a block diagram showing the frequency detector according to another preferred embodiment of the present invention. The apparatus of Figure 7 comprises a sampling portion 30, a phase difference detector 32, and a frequency offset signal generator 40 employing formula (32) regarding the QPSK method. Frequency offset signal generator 40 comprises sign generators 41, 42, 45 and 46, a subtractor 43, adders 44 and 49, and multipliers 47 and 48. These constituent components are connected according to the above formula (32). In Figure 7, second frequency offset signal generator 38 (Figure 4) may be additionally connected to the output of frequency offset signal generator 40. The operation of the apparatus shown in Figure 7 will now be described. Sampling portion 30 outputs signal R k sampled at the sampling frequency which is the same as the symbol rate. Phase difference detector 32 performs the same operation as that of the block illustrated in Figure 4. Phase detection signal U k output from phase difference detector 32 is applied to first and second sign generators 41 and 42. Also, real number component signal C k is applied to first multiplier 47 and imaginary number component signal S k is applied to second multiplier 48. First sign generator 41 outputs the value of the sign function with respect to the difference (C k -S k ) between the real and imaginary number components of phase detection signal U k as the first sign signal. Second sign generator 42 outputs the value of the sign function with respect to the sum (C k +S k ) of the real and imaginary number components of phase detection signal U k as the second sign signal. Subtractor 43 to which the first and second sign signals are applied subtracts the second sign signal from the first sign signal and outputs the result. First adder 44 receives the first and second sign signals, adds them together and outputs the result. Third sign generator 45 outputs the result value of the sign function with respect to the output of subtractor 43 as the third sign signal. Fourth sign generator 46 outputs the result value of the sign function with respect to the output of adder 44 as the fourth sign signal. First multiplier 47 multiplies the third sign signal by real number component signal C k and outputs the result. Second multiplier 48 multiplies the fourth sign signal by imaginary component signal S k and outputs the result. Second adder 49 receives the outputs of multipliers 47 and 48 and generates first frequency offset signal sin (ΔωT). Figure 8 is a block diagram showing the frequency detector according to still another preferred embodiment of the present invention. The apparatus of Figure 8 comprises a sampling portion 30, a phase difference detector 32, and a frequency offset signal generator 50 employing formula (32) regarding the QPSK method. In frequency offset signal generator 50, a control signal generator 51 receives phase detection signal U k , signal inverters 52 and 53 invert the signal component C k or S k output from phase difference detector 32 and outputs the inverted signals, and multiplexer 54 selects the signals input according to the selection control signal (SEL) generated by control signal generator 51 and outputs the result as first frequency offset signal sin (ΔωT). In Figure 8, second frequency offset signal generator 38 (Figure 4) may be additionally connected to the output of frequency offset signal generator 50. The operation of the apparatus shown in Figure 8 will be described. When sampling portion 30 outputs signal R k sampled at the sampling frequency which is the same as the symbol rate, phase difference detector 32 performs the same operation as that of the block illustrated in Figure 4. Real number component signal C k of phase detection signal U k is inverted by first signal inverter 52 and imaginary number component signal S k is inverted by second signal inverter 53. Control signal generator 51 is constructed for generating the selection control signal (SEL) according to the above formula (32). That is, the selection control signal is the sign signal of two bits representing the logical product of sgn (C k -S k ) and sgn (C k +S k ). Multiplexer 54 selects one signal among four signals S k , -S k , C k and -C k by the selection control signal and outputs the selected signal. The signal output from multiplexer 54 is first frequency offset signal sin (ΔωT), which is S k if C k -S k > 0 and C k +S k > 0, is -C k if C k -S k < 0 and C k +S k > 0, is -S k if C k -S k < 0 and C k +S k < 0, and is C k if C k -S k > 0 and C k +S k < 0. The description of the signal processor for generating frequency offset signal ΔωT in the preferred embodiments of Figures 7 and 8 has been omitted. However, those skilled in the art can construct such an apparatus which receives the frequency offset signal expressed as sin (ΔωT) to generate (ΔωT), using a ROM table. The present invention can be used for the automatic frequency control of a modem using all types of MPSK modulation. As described above, the apparatus according to present invention detects the frequency offset generated by the distance between the frequency of a carrier wave and the local oscillation frequency of a receiver or by the Doppler shift, using the transmission phase information and the phase difference information between the currently sampled signal and the immediately preceding sampled signal, thereby providing the effect of automatic frequency control in a MPSK communication method. Also, according to the present invention, the frequency offset is detected using the signal sampled at a symbol rate, thereby resulting in the effect of cost reduction in the hardware for embodying the present invention.

Claims: An automatic frequency control method, in which a MPSK-modulated signal is received, a frequency offset signal is generated using the phase information of the modulated signal, and the oscillation frequency of a receiver is controlled on the basis of the frequency offset signal, comprising the steps of: sampling the received signal at a predetermined sampling frequency; generating a phase detection signal having the phase difference information between the currently sampled signal and the immediately preceding sampled signal; determining a reference phase which is the closest to the phase difference information of the phase detection signal among the reference phase values of the M-ary phase-shift keying of said modulated signal, as transmission phase information of said received signal; and generating a frequency offset signal corresponding to the phase difference between said phase difference information and said determined transmission phase information. An automatic frequency control method as claimed in claim 1, wherein said phase detection signal generating step comprises a normalization step where a first phase detection signal U k is generated according to U k = R k R * k -1 R k x R k -1 = e j ( k +ΔωT) ≡ C k + jS k by using a currently sampled signal R k and an immediately preceding sampled signal R k-1 where  k represents the difference between phase information transmitted from the current symbol period and the phase information transmitted from the immediately preceding symbol period, and ΔωT is a frequency offset signal. An automatic frequency control method as claimed in claim 1, wherein said phase detection signal generating step comprises a complex conjugation operation step where a second phase detection signal U' k is generated according to U' k = R k R* k-1 by using a currently sampled signal R k and an immediately preceding sampled signal R k-1 . An automatic frequency control method as claimed in claim 1 or claim 2, wherein said frequency offset signal generating step comprises a first frequency offset signal generation step where a first frequency offset signal sin (ΔωT) is generated according to sin (ΔωT) = Im[U k e -jψi ] using said reference phase ψ i and said phase detection signal U k . An automatic frequency control method as claimed in claim 4, wherein said frequency offset signal generating step further comprises a second frequency offset signal generation step where a second frequency offset signal ΔωT is generated using the phase of said first frequency offset signal sin (ΔωT). An automatic frequency control method as claimed in claim 5, wherein, in said transmission phase information determination step, when said phase difference information ∠U k of phase detection signal U k includes a second frequency offset signal ΔωT between -π/M and π/M determined according to the number M of the reference phase of the M-ary phase-shift keying, and when said phase difference information ∠U k , second frequency offset signal ΔωT and transmission phase information  k satisfy ΔωT = ∠U k -  k , a reference phase ψ i is determined as the reference phase which is the closest to said phase difference information. An automatic frequency control apparatus, in which a MPSK-modulated signal is received, a frequency offset signal is generated using the phase information of the modulated signal, and the oscillation frequency of a receiver is controlled on the basis of the frequency offset signal, comprising: sampling means (30) sampling the received signal at a predetermined sampling frequency and outputting the sampled signal (R k ); phase difference detection means (32) receiving the currently sampled signal (R k ) from said sampling means (30) and for generating a phase detection signal (U k ) having phase difference information between the currently sampled signal (R k ) and the immediately preceding sampled signal (R k-1 ) stored therein; and means (34, 36) receiving said phase detection signal (U k ) to determine a reference phase which is the closest to said phase difference information of said phase detection signal (U k ) among the reference phase values of the M-ary phase-shift keying of said modulated signal, as transmission phase information of said received signal, and generating a frequency offset signal determined according to the phase difference between said phase difference information and said transmission phase information. An automatic frequency control apparatus as claimed in claim 7, wherein said phase difference detection means (32) comprises normalization means for receiving the currently sampled signal (R k ) and the immediately preceding sampled signal (R k-1 ) and generating said phase detection signal (U k ). An automatic frequency control apparatus as claimed in claim 7, wherein said phase difference detection means (32) comprises a complex conjugation operator for receiving the currently sampled signal (R k ) and the immediately preceding sampled signal (R k-1 ) and generating said phase detection signal (U k ). An automatic frequency control apparatus as claimed in any one of claims 7 to 9, wherein said frequency offset signal generation means (34, 36) comprises: determination means (34) for storing a plurality of critical values determined by said reference phase values, receiving said phase detection signal (U k ) and outputting the reference phase value determined by said phase difference information of said phase detection signal (U k ) and said plurality of critical values; and first frequency offset signal generator (36) for receiving said transmission phase information and said phase detection signal (U k ) and generating first frequency offset signal which is an imaginary number component generated through the complex conjugation operation between the received transmission phase information and phase detection signal. An automatic frequency control apparatus as claimed in claim 10, wherein said first frequency offset generator (36) receives said reference phase signal e -jψi and said phase detection signal U k from said determination means and generates a first frequency offset signal sin (ΔωT) determined according to sin (ΔωT) = Im[U k e -jψi ] = S k cos ψ i -C k sin ψ i . An automatic frequency control apparatus as claimed in claim 11, further comprising a second frequency offset signal generator (38) for receiving said first frequency offset signal ( sin (ΔωT)) and generating a second frequency offset signal (ΔωT) through an arc sine operation. An automatic frequency control apparatus as claimed in claim 12, wherein, in said determination means, when said phase difference information ∠U k of said phase detection signal U k includes a second frequency offset signal ΔωT between -π/M and π/M determined according to the number M of reference phase of phase-shift keying, and when said phase difference information ∠U k , second frequency offset signal ΔωT and transmission phase information  k satisfy ΔωT = ∠U k -  k , a reference phase ψ i is determined as the reference phase which is the closest to said phase difference information. An automatic frequency control apparatus as claimed in any one of claims 7 to 13, wherein said sampling means (30) samples said received signal at the sampling frequency equal to the symbol rate. An automatic frequency control apparatus as claimed in claim 7, wherein said frequency offset signal generator (40) comprises: a first sign generator (41) for receiving the QPSK-modulated phase detection signal and generating a first sign signal which is the value of a sign function with respect to the difference obtained by subtracting a second component signal constituting said phase detection signal from first component signal constituting said phase detection signal; a second sign generator (42) for generating a second sign signal which is the value of a sign function with respect to the sum of said first and second component signals; a subtractor (43) for subtracting said second sign signal from said first sign signal; a first adder (44) for summing said first and second sign signals; a third sign generator (45) for outputting a third sign signal which is the sign function value of the signal output from said subtractor (43) ; a fourth sign generator (46) for outputting a fourth sign signal which is the sign function value of the signal output from said adder (44) ; a first multiplier (47) for multiplying said first component signal by the output signal of said third sign generator (45) ; a second multiplier (48) for multiplying said second component signal by the output signal of said fourth sign generator (46); and a second adder (49) for summing the output signals of said first and second multipliers (47, 48) and generating said first frequency offset signal. An automatic frequency control apparatus as claimed in claim 7, wherein said frequency offset signal generator (50) comprises: a control signal generator (51) for receiving the QPSK-modulated phase detection signal and generating a selection control signal determined according to a first sign function value with respect to the difference obtained by subtracting the second component signal constituting said phase detection signal from the first component signal constituting said phase detection signal and a second sign function value with respect to the sum of said first and second component signals, said control signal generator (51) generating said selection control signal so that the first component signal is selected as said first frequency offset signal if the first and second sign function values are greater than zero, the inverted signal of the first component signal is selected as said first frequency offset signal if the first sign function value is less than zero and the second sign function value is greater than zero, the inverted signal of the second component signal is selected as said first frequency offset signal if the first and second sign function values are less than zero, and the second component signal is selected as said first frequency offset signal if the first sign function value is greater than zero and the second sign function value is less than zero; a first signal inverter (52) for inverting and outputting said first component signal; a second signal inverter (53) for inverting and outputting said second component signal; and selection means (54) comprising four signal inputs connected for receiving said first and second component signals and the output signals of said first and second signal inverters (52, 53), for selecting one of said four input signals according to said selection control signal and outputting said selected signal as said first frequency offset signal. An automatic frequency control apparatus as claimed in claim 15 or claim 16, further comprising a second frequency offset signal generator for receiving said first frequency offset signal and generating a second frequency offset signal through an arc sign operation. An automatic frequency control apparatus as claimed in any one of claims 15 to 17, wherein said first component signal is a real number component of said phase detection signal and said second component signal is an imaginary number component of said phase detection signal. An automatic frequency control apparatus as claimed in any one of claims 15 to 18, wherein said sampling means samples said received signal at the sampling frequency equal to the symbol rate.

Inventors: CHOI YANG-SEOK, CHOI, YANG-SEOK
Applicants: SAMSUNG ELECTRONICS CO LTD, SAMSUNG ELECTRONICS CO., LTD.
Application year: 1995
Main classification: H03J7
IPC codes: H03J7, A45D20, H04L27