Variable speed control apparatus for induction motor

Abstract: A variable speed regulating apparatus is comprised of a three-phase/two-phase transformer and a vector rotator for obtaining an induced voltage vector; an operational circuit which obtains an M axis component (E M ) and a T axis component (E T ) of said induced voltage vector based on a magnetic flux position; and a primary angular frequency operating means which operates a primary angular frequency command value (ω₁*) by correcting a primary angular frequency (ω₁) obtained by dividing the absolute value of said induced voltage vector or its T axis component (E T ) by a magnetic flux equivalent, with the output of a regulating means (52) which regulates said M axis component (E M ) to zero, and feeds the primary angular frequency command value (ω₁*) to an integrator. The output of the regulating means (52) is multiplied by a variable (F(x)) so as to substantially reduce the proportional gain of the regulating means (52) in a field weakening range.

Description: The present invention relates to a variable speed control apparatus which controls a motor by the transvector control method without using a speed sensor. Transvector control systems, which control a motor, without using a speed sensor, by operating a primary angular frequency for detecting a position of a magnetic flux axis based on an induced voltage of the motor, are disclosed in JP-A-64-8896 and JP-A-01-198292, the latter document disclosing an apparatus according to the pre-characterizing portion of claim 1. Fig. 15 is a block diagram of a variable speed control apparatus almost similar to the variable speed control apparatuses described in the above mentioned documents. Fig. 16 is a block circuit diagram showing a primary angular frequency operating means 200' of Fig. 15. A vector diagram of an induced voltage is shown in Fig. 3 and will be explained later. Here, the operation of the primary angular frequency operating means 200' according to the prior art is briefly explained. In Fig. 16, an absolute value operating circuit 53 calculates the absolute value | E T | of a T axis component E T (T axis induced voltage) of an induced voltage vector E. Here, an M axis is an axis of a rotational coordinate system lying in a direction of magnetic flux, and the T axis is an axis perpendicular to the M axis. The absolute value | E T | is fed to a divider 28. The divider 28 calculates a primary angular frequency by dividing the absolute value | E T | by a magnetic flux command value  2 *. As Fig. 3 indicates, an M axis component E M (M axis induced voltage) of the induced voltage vector E has a non-zero value when a phase angle command value *, obtained from an integrator 102 of Fig. 15, deviates from an actual phase angle value  of a motor. To avoid this, the primary angular frequency operating means 200' conducts a control operation which always controls E M at zero by inputting E M to a regulating means 52, and subtracting in the adder 55 the output of the regulating means 52 from the output of the divider 28. By this control operation, the absolute value of a primary angular frequency | ω 1 * | is corrected so that the phase angle command value * may coincide with the actual phase angle value e. A P controller which conducts proportional operation or Pl controller which conducts proportional plus integral operation is used for the regulating means 52. The derivation of the primary angular frequency command value ω 1 * is completed by detecting the sign of E T in a polarity detecting circuit 54, and adding the detected sign to | ω 1 * | in a multiplier 51. The thus obtained ω 1 * is converted, in integrator 102 of Fig. 15 as a magnetic flux position operating means, to the phase angle command value *, which is used for voltage and current vector rotating operations in vector rotators 11 and 24, and in a coordinate transformer 8. An adder 202 calculates an estimated speed value ω r by subtracting from ω 1 * a slip frequency command value ω S * calculated in a slip frequency operator 101. (Though a ^ symbol is added to estimated values throughout the drawing figures, this symbol is omitted in the specification.) The estimated speed value ω r is used for speed control of an induction motor 2. In Fig. 15, a primary current of the induction motor 2 is detected by a current detector 203, and resolved in a three-pase/two-phase transformer 12 into two-phase values i α , i β . which are further transformed in vector rotator 11 to an actual magnetizing current value l M and an actual torque current value I T on the above described rotational coordinate system (M-T coordinates) defined with respect to the axis of magnetic flux. A magnetic flux regulator 4 generates a magnetizing current command value l M * based on the magnetic flux command value  2 *, and a speed regulator 5 generates a torque current command value I T * from a speed command value ω r * and the estimated speed value ω r . A current regulator 6 obtains an M component V M * (magnetizing voltage command value) and a T component V T * (torque voltage command value) of a primary voltage command value from the command values l M *, l T * and the actual values l M , l T . The voltage command values V M *, V T * are transformed though a coordinate transforming circuit 8 to two-phase values V α *, V β * which are further converted in a pulse generating circuit 9 to drive pulses for driving a PWM inverter 1. A voltage detector 20, a three-pase/two-phase transformer 21, the vector rotator 24 and induced voltage operating circuit 22 are used for detecting the induced voltage vector E of the induction motor 2, and for obtaining its components E M , E T . EP-A-0 105 511 discloses a control method for induction motors wherein the primary frequency is controlled so that M axis component E M (referred to as d-axis component e d in the document) is kept at zero. There is no disclosure about control in the field-weakening region in this document. In the prior art described above, the output of the regulating means 52, which conducts proportional or proportional plus integral operation for regulating the M axis induced voltage E M at zero, is directly added to the absolute value of the primary angular frequency | ω 1 * |. Because of this, the proportional gain of the regulating means 52 becomes too large relative to the intensity of the magnetic flux in a field-weakening region in which the magnetic flux intensity of the induction motor 2 is weakened. This relatively too large proportional gain of the regulating means 52 affects the primary angular frequency command value ω 1 *, and is hazardous for a stable and smooth drive of the induction motor 2. Since the induced voltage becomes small in a low speed driving range, decrease of an S/N ratio of the detected voltage signal and significant digits of the detected data causes a large operational error of the primary angular frequency command value obtained by dividing the T axis induced voltage value E T by the magnetic flux intensity. Since the magnetic flux command value  2 * is used as an equivalent of the magnetic flux intensity in the above described dividing, an error is caused in the primary angular frequency command value ω 1 * when the command value  2 * deviates from the actual value of the magnetic flux which may vary greatly depending on the driving condition of the induction motor. In view of the foregoing, a first object of the present invention is to provide a variable speed control apparatus for an induction motor which facilitates driving the induction motor stably and smoothly even in the field weakening range. A second object of the present invention is to provide a variable speed control apparatus for an induction motor which reduces the operational error of the primary angular frequency command value in a low speed driving range. And, a third object of the present invention is to provide a variable speed control apparatus for an induction motor which reduces the operational error of a magnetic flux equivalent which is necessary for calculating the primary angular frequency command value from the induced voltage. These objects are achieved with an apparatus as claimed in Claim 1. In a variable speed control apparatus for an induction motor according to the present invention, the primary angular frequency operating means is provided with a multiplier which multiplies the output of the first regulating means by a variable which varies in response to the magnetic flux intensity. This first regulating means regulates the M axis induced voltage at zero. The primary angular frequency operating means corrects the primary angular frequency command value by the output of the multiplier. The primary angular frequency operating means is preferably provided with an adder which adds the output of second regulating means and the output of the primary angular frequency operating means described above. The second regulating means regulates (by proportional or proportional plus integral operation) the deviation between the command value of the torque current or the torque and an actual value to zero. A variable speed control apparatus for an induction motor is preferably provided with a magnetic flux operating means which calculates first and second values. The first value is obtained by adding the result of multiplication of a reference frequency value by the magnetic flux command value and the absolute value of the induced voltage or the T axis induced voltage. The second value is obtained by adding the reference frequency value and the absolute value of the primary angular frequency command value. The magnetic flux operating means divides the first value by the second value and obtains a calculated magnetic flux value by filtering the result of the division through a low path filter. The calculated magnetic flux value is used as a magnetic flux equivalent necessary for the input of the primary angular frequency operating means and the magnetic flux regulator. The relation among the M axis induced voltage E M , the T axis induced voltage E T , the primary angular frequency ω 1 and the secondary magnetic flux  2 is expressed by equation 1. In equation 1, sgn(E T ) indicates the polarity (+, -) of E T . ω 1 = sgn (E T )(|E|/ 2 ) = sgn(E T ) [√(E M 2 +E T 2 )]/ 2 Fig. 2 is a vector diagram illustrating the relation among the voltage, the current and the magnetic flux of the induction motor. Since the M axis is directed along the direction of the secondary magnetic flux  2 by transvector control, the M axis induced voltage E M usually becomes zero as equation 2 indicates: E M = 0 And, when equation 2 holds, equation 3 also holds. ω 1 = E T / 2 Transvector control of an induction motor can be realized, without using data from a speed detector of the induction motor, by generating the phase angle command value * by integrating the primary angular frequency command value ω 1 * calculated from equation 1 or 3, and by obtaining a position of the magnetic flux. Fig. 3 is a voltage vector diagram of the induction motor when the above described phase angle command value * deviates from the actual phase angle  of the motor. The rotational coordinate system based on the phase angle command value * is expressed by the M-T coordinates, and the rotational coordinate system based on the actual phase angle  is expressed by the M'-T' coordinates. Since the direction of the induced voltage vector E coincides with the direction of the T' axis, the M axis induced voltage E M has a value corresponding to the error of the phase angle when * deviates from . As Fig. 3 indicates, E M >0 when * is in advance of e, and E M <0 when * is in retard of . Therefore, by disposing the first regulating means to which E M is input and by subtracting the output of the first regulating means from the primary angular frequency command value, the absolute value |ω 1 *| of the primary angular frequency command value is corrected so that E M may be always zero. By utilizing the fact that the sign of the primary angular frequency ω 1 and the T axis induced voltage E T coincide with each other, the primary angular frequency command value ω 1 * is obtained by adding the sign of E T to the absolute value of the primary angular frequency. As a result of the correction described earlier which regulates E M at zero, a phase angle of the motor fed to the magnetic flux position operating means coincides with the actual phase angle of the motor. At that time, the primary angular frequency command value ω 1 * is calculated from equation 4. In equation 4, G em is a transfer function of the first regulating means to which E M is input. The primary angular frequency operating means 200' of Fig. 16 according to the prior art uses the magnetic flux command value  2 * in place of the actual secondary magnetic flux value  2 of equation 4 which can not be detected directly. ω 1 * = sgn (E T )[( E T / 2 ) - G em E M ] In the field weakening range in which the magnetic flux intensity becomes small, since the magnetizing current becomes relatively small with respect to the torque current and the phase angle of the magnetic flux tends to change, it is necessary to reduce the gain of the first regulating means for driving the motor stably and smoothly. Based on this, the present invention realizes stable and smooth driving of the motor by reducing the gain of the first regulating means in the field weakening range by multiplying the output of the first regulating means by a variable F(x) which varies in response to the intensity of the magnetic flux. In practice, this correction is conducted by multiplying E M in the correction term of equation 4 by the variable F(x). Then, ω 1 * is expressed by equation 5. ω 1 * = sgn (E T )[( E T / 2 ) - G em F(x)E M ] Since the induced voltage is small in the low speed driving range, a voltage detection error causes a relatively large error of the primary angular frequency command value ω 1 *. Based on the fact that the speed of the motor is varied by the variation of the torque current or torque, the speed control apparatus preferably corrects the primary angular frequency command value ω 1 * at low speed by adding the output of the second regulating means, which regulates the torque current or the torque at the torque current command value or the torque command value, to the primary angular frequency command value. The error of ω 1 * is also reduced by the present invention, since the current value has more significant digits than the voltage value in the low speed range. Thus, the present variable speed control apparatus realizes stable and smooth driving of the induction motor. Though the variable speed control apparatus according to the prior art uses the magnetic flux command value  2 * as an equivalent of the magnetic flux necessary for obtaining the primary angular frequency command value ω 1 *, an error is caused when the secondary magnetic flux of the motor varies greatly. By setting E M =0, and by modifying equation 1, equation 6 is obtained.  2 = E T /ω 1 If equation 6 is used as it is for calculating the secondary magnetic flux  2 , it is difficult to calculate the secondary magnetic flux  2 , since both E T and ω 1 are very small in the low speed driving range. To avoid this, the variable speed control apparatus introduces a reference frequency value a of finite value and obtains the calculated magnetic flux value  2 .  2 = (α 2 *+ E T )/(α+ ω 1 * ) In the low speed range in which E T is very small, when the primary angular frequency command value ω 1 * is sufficiently smaller than the reference frequency value α,  2 =  2 * from equation 7, and when the primary angular frequency command value ω 1 * is sufficiently larger than the reference frequency value α,  2 = | E T /ω 1 * |. Thus, by using equation 7 in place of equation 6, the magnetic flux value is calculated with a small error over the entire speed range. However, ω 1 * and  2 can not be calculated simultaneously, since equation 7 requires the primary angular frequency command value ω 1 * for calculating the calculated magnetic flux value  2 , and since equation 5 requires the calculated magnetic flux value  2 for calculating the primary angular frequency command value ω 1 *. Though the primary angular frequency ω 1 instantaneously changes, the secondary magnetic flux  2 changes slowly at a speed determined by a secondary time constant of the motor. Based on this fact, the variable speed control apparatus solves the above described problem by providing a magnetic flux operating means, on its output side, with a low path filter as described later so that  2 may change slowly at a speed corresponding to the secondary time constant of the motor. Thus, ω 1 * and  2 are calculated separately in time series with the above described arrangement which instantaneously calculates ω 1 * based on  2 which changes slowly. Now the present invention will be described in detail hereinafter with reference to the drawings which illustrate preferred embodiments of the present invention only and in which: Fig. 1 is a block diagram of a variable speed control apparatus using the present invention; Fig. 2 is a vector diagram illustrating the relation among the voltage, the current and the magnetic flux of the induction motor; Fig. 3 is a vector diagram illustrating the components of the induced voltage etc.; Fig. 4 is a block diagram of the first invention embodiment of a primary angular frequency operating means of Fig. 1; Fig. 5 is a block diagram of the second invention embodiment of the primary angular frequency operating means of Fig. 1; Fig. 6 is a block diagram of the third invention embodiment of the primary angular frequency operating means of Fig. 1; Fig. 7 is a block diagram of the fourth invention embodiment of the primary angular frequency operating means of Fig. 1; Fig. 8 is a block diagram of the fifth invention embodiment of the primary angular frequency operating means of Fig. 1; Fig. 9 is a block diagram of the sixth invention embodiment of the primary angular frequency operating means of Fig. 1; Fig. 10 is a block diagram showing the correlation between the primary angular frequency operating means and the magnetic flux operating means of Fig. 1; Fig. 11 is a block diagram showing the first embodiment of the magnetic flux operating means of Fig. 1; Fig. 12 is a block diagram showing the second embodiment of the magnetic flux operating means of Fig. 1; Fig.13 is a block diagram showing the third embodiment of the magnetic flux operating means of Fig. 1; Fig. 14 is a block diagram showing the fourth embodiment of the magnetic flux operating means of Fig. 1; Fig. 15 is a block diagram of a variable speed control apparatus according to the prior art; and Fig. 16 is a block diagram of a primary angular frequency operating means according to the prior art. Fig. 1 is a block diagram of a variable speed control apparatus using the present invention. The basic structure of the variable speed control apparatus of Fig. 1 is common to all the embodiments using the present invention. In Fig. 1, a primary current of a motor 2 is resolved in a three-phase/two-phase transformer 12 into two-phase current components i α , i β on the stator coordinate system, and is further transformed in a vector rotator 11 to an actual magnetizing current value l M and an actual torque current value IT on the M-T coordinate system defined with respect to the axis of magnetic flux. A magnetic flux regulator 4 generates a magnetizing current command value l M * from a deviation between a magnetic flux command value  2 * and a calculated magnetic flux  2 . A speed regulator 5 generates a torque current command value I T * from a speed command value ω r * and the estimated speed value ω r . A current regulator 6 obtains an M component V M * and a T component V T * of a primary voltage command value from the command values l M *, l T * and the actual values l M , l T . The voltage command values V M *, V T * are transformed though a coordinate transforming circuit 8 to two-phase values V α *, V β * which are further converted in a pulse generating circuit 9 to drive pulses for driving a PWM inverter 1. The estimated speed value ω r described above is obtained by subtracting a slip frequency command value ω S * from a primary angular frequency command value ω 1 *. A slip frequency operator 101 outputs the slip frequency command value ω S * based on the actual torque current value I T and the calculated magnetic flux  2 . The calculated magnetic flux  2 is output from a magnetic flux operating means 201. In Fig. 1, a voltage detector 20 and a three-phase/two-phase transformer 21, a vector rotator 24 and an induced voltage operating circuit 22 constitute a means for obtaining a induced voltage vector of the induction motor 2. Induced voltage operating circuit 22 constitutes a voltage component operating means for obtaining an M axis component E M of the induced voltage vector E of the induction motor 2, and for obtaining a T axis components E T perpendicular to E M . Fig. 4 is a block diagram of the first invention embodiment of a primary angular frequency operating means 200 of Fig. 1. The principle for obtaining the primary angular frequency command value is similar to that described in JP-A-64-8896. In Fig. 4, an absolute value operating circuit 53 calculates the absolute value of the T axis induced voltage E T . The absolute value of E T is fed to a divider 28. The divider 28 divides the input absolute value of E T by the calculated magnetic flux  2 according to equation 3, and outputs the absolute value | ω 1 | of the primary angular frequency. As shown in Fig. 4, primary angular frequency operating means 200 is provided with a regulating means 52 (the first regulating means) to which E M is input, and a multiplier 50 which multiplies the output of the regulating means 52 by a variable F(x) which changes in response to the intensity of the secondary magnetic flux. An adder 55 subtracts the output of the multiplier 50 from the absolute value of the primary angular frequency | ω 1 | output from the divider 28. By this operation, the primary angular frequency is corrected so as to regulate E M at zero based on the principle explained earlier. Since the gain of the regulating means 52 is adjusted by the multiplier 50 so as to be proportional to the variable F(x), stable and smooth driving of the induction motor 2 is facilitated. The magnetic flux command value  2 * or the magnetizing current command value l M * which corresponds to the magnetic flux intensity may be used as the variable F(x). Since the sign of the primary angular frequency coincides with the sign of the T axis induced voltage E T , the primary angular frequency command value ω 1 * having the same polarity with the T axis induced voltage E T is obtained by multiplying in a multiplier 51 the output of the adder 55 by the sign of E T detected by a polarity detecting circuit 54. By lowering the gain of the regulating means 52, stable and smooth driving of the induction motor 2 is facilitated in the field weakening range even when the magnetizing current value has less significant digits than the torque current and the phase angle of the magnetic flux trends to change. Fig. 5 is a block diagram of the second invention embodiment of the primary angular frequency operating means 200 of Fig. 1. The primary angular frequency operating means of Fig. 5 is different from the first embodiment of Fig. 4 in that the M axis induced voltage E M and the T axis induced voltage E T are input to the absolute value operating circuit 53, and the circuit 53 outputs the absolute value of the induced voltage vector | E |(= √(E M 2 +E T 2 )). In this embodiment, the primary angular frequency command value ω 1 * is calculated by equation 1. Fig. 6 is a block diagram of the third invention embodiment of the primary angular frequency operating means 200 of Fig. 1. The primary angular frequency operating means of Fig. 6 is different from the first embodiment of Fig. 4 in that the T axis induced voltage E T is input directly to the divider 28, and that the multiplier 51 is connected between the multiplier 50 and the adder 55. The multiplier 51 inputs the output of the multiplier 50 to the adder 55 after providing the output of the multiplier 50 with the sign of E T detected by the polarity detecting circuit 54. Fig. 7 is a block diagram of the fourth invention embodiment of the primary angular frequency operating means 200 of Fig. 1. The primary angular frequency operating means of Fig. 7 includes the arrangement of the first embodiment of Fig. 4 provided with an adder 62 which adds a frequency correction value to the primary angular frequency command value ω 1 *. This frequency correction value is obtained by calculating a deviation by subtracting in an adder 60 the actual torque current value I T or an actual torque value (or a calculated torque value) τ from the torque current command value l T * or a torque command value τ*, and by conducting in a regulating means 61 (the second regulating means) proportional or proportional plus integral operation so as to regulate the deviation at zero. This embodiment reduces the detection error or operational error and facilitates stable and smooth driving of the induction motor even in the low speed driving range in which the induced voltage becomes small, by correcting the primary angular frequency command value ω 1 * with the torque current value etc. which has more significant digits than the induced voltage. Fig. 8 is a block diagram of the fifth invention embodiment of the primary angular frequency operating means 200 of Fig. 1. The primary angular frequency operating means of Fig. 8 includes the arrangement of the second embodiment of Fig. 5 provided with an adder 62 which adds a frequency correction value to the primary angular frequency command value ω 1 *. The primary angular frequency is corrected in this embodiment in the same way as in the fourth embodiment of Fig. 7. Fig. 9 is a block diagram of the sixth invention embodiment of the primary angular frequency operating means 200 of Fig. 1. The primary angular frequency operating means of Fig. 9 includes the arrangement of the third embodiment of Fig. 6 provided with an adder 62 which adds a frequency correction value to the primary angular frequency command value ω 1 *. Also in this embodiment, the primary angular frequency is corrected in the same way as in the fourth embodiment of Fig. 7. Fig. 10 is a block diagram showing the correlation between the primary angular frequency operating means 200 and the magnetic flux operating means 201. The magnetic flux operating means 201 obtains the magnetic flux value  2 by calculating equation 7 using the input primary angular frequency command value ω 1 *, T axis induced voltage E T , and magnetic flux command value  2 *. The calculated magnetic flux value  2 is filtered through a low path filter 76 and is converted to the final calculated value of magnetic flux  2 which changes at the time constant of the induction motor. This final calculated magnetic flux value  2 is input to the primary angular frequency operating means 200 and used in the dividing operation in the divider 28 in all the above described embodiments. Fig. 11 is a block diagram showing the first embodiment of the magnetic flux operating means 201. In Fig. 11, an absolute value operating circuit 70 calculates the absolute value of the T axis induced voltage E T , and a multiplier 72 multiplies the magnetic flux command value  2 * by a reference frequency constant α. An adder 73 adds the output of the absolute value operating circuit 70 and the output of the multiplier 72. An absolute value operating circuit 71 calculates the absolute value of the primary angular frequency command value ω 1 *, and an adder 74 adds the reference frequency constant α and the output of the absolute value operating circuit 71. A divider 75 divides the output of the adder 73 by the output of the adder 74. Finally the calculated magnetic flux value  2 is obtained by filtering the output of the divider 75 through a low path filter 76. The calculated magnetic flux value  2 thus obtained is used as a magnetic flux equivalent in the above described embodiments of the primary angular frequency operating means. This embodiment facilitates obtaining from equation 7 the calculated magnetic flux value  2 with a small error in either case whether the primary angular frequency command value ω 1 * is sufficiently smaller or larger than the reference frequency value α. Since the calculated magnetic flux value  2 obtained through the low path filter 76 changes more slowly than the primary angular frequency command value ω 1 *,  2 and ω 1 * can be separately calculated in time series. Due to this, since  2 and ω 1 * can substantially be calculated simultaneously, equation 5 can be calculated with no troubles. Fig. 12 is a block diagram showing the second embodiment of the magnetic flux operating means 201. This embodiment is different from the first embodiment of Fig. 11 in that the absolute value operating circuit 70 calculates the absolute value of the induced voltage vector E instead of the T axis induced voltage E T . Explanation on the other arrangement, which is the same with the first embodiment of Fig. 11, will be omitted. Fig. 13 is a block diagram showing the third embodiment of the magnetic flux operating means. In this embodiment, the magnetic flux command value  2 * in the first embodiment of Fig. 11 is replaced with the magnetizing current command value l M * filtered through a low path filter 77. Explanation on the other arrangement, which is the same with the first embodiment of Fig. 11, will be omitted. Fig. 14 is a block diagram showing the fourth embodiment of the magnetic flux operating means. In this embodiment, the magnetic flux command value  2 * in the second embodiment of Fig. 12 is replaced with the magnetizing current command value l M * filtered through a low path filter 77. Explanation on the other arrangement, which is the same with the second embodiment of Fig. 12, will be omitted. According to the present invention, stable and smooth driving of an induction motor is facilitated by lowering the gain of the first regulating means of the primary angular frequency operating means in the field weakening region in which the phase of the magnetic flux tends to vary. As explained above, the output of the second regulating means which regulates the torque current or the torque at the command value is added to the output of the primary angular frequency operating means. By this measure, the primary angular frequency command value can be obtained with a small error by correcting the primary angular frequency value with the current value or the torque value which has more significant digits than the voltage value and is detectable more precisely even in the low speed driving range in which detected voltage is low and the operational error of the induced voltage becomes relatively large. Since the magnetic flux is obtained on the basis of the induced voltage in the high or middle speed driving range of the motor and on the basis of the magnetic flux command value in the low speed driving range in which the induced voltage detection error becomes large, the magnetic flux value can be calculated accurately over the entire speed range of the motor. And, due to this, the primary angular frequency command value can also be calculated accurately.

Claims: A variable speed control apparatus for an induction motor (2) comprising: power converting means (1) capable of controlling magnitude, frequency and phase of the output voltage thereof, for supplying power to said induction motor; magnetic flux position operating means (102) for establishing the position of magnetic flux for controlling a magnetizing current component (l M ) and a torque current component (l T ) independently, whereby to control at least the torque of said induction motor; means (11, 12) responsive to said magnetic flux position operating means (102) for resolving a primary current of said induction motor into a magnetizing current component (l M ) and a torque current component (l T ); means (21, 22, 24) for obtaining an induced voltage vector (E) of said induction motor; component operating means (22) responsive to said magnetic flux position operating means (102) for obtaining a magnetic flux axis component (E M ) and a torque axis component (ET) of said induced voltage vector based on said magnetic flux position (*); and primary angular frequency operating means (200), including a first regulating means (52) for regulating said magnetic flux component (E M ) at zero thereby to correct a primary angular frequency (ω 1 ), obtained by dividing the absolute value of said torque axis component (E T ) of said induced voltage vector or the absolute value of said induced voltage vector (E) by a magnetic flux equivalent ( 2 ) , with the output of said first regulating means (52) and for obtaining a primary angular frequency command value (ω 1 *) fed to said magnetic flux position operating means (102), characterized in that said primary angular frequency operating means (200) further comprises means (50) for reducing the gain of said first regulating means (52) in a field weakening range. The apparatus as claimed in claim 1, further comprising second regulating means (61) for regulating said torque current component (I T ) or said torque (T) at a command value (l T *, τ*) thereof, and adding means (62) for adding the output of said second regulating means (61) to the output of said primary angular frequency operating means (200). The apparatus as claimed in claim 1 or 2, further comprising a magnetic flux operating means (201) for obtaining a first value by adding a value, obtained by multiplying a magnetic flux command value ( 2 *) by a reference frequency value (α), and the absolute value ( | E | ; | E T | ) of said induced voltage vector or its torque axis component, for obtaining a second value by adding said reference frequency value (α) and the absolute value ( | ω 1 * | ) of said primary angular frequency command value, and for dividing said first value by said second value, said magnetic flux operating means further comprising a low path filter means (76) for filtering the result of said division and for feeding said filtered result to said primary angular frequency operating means (200) as said magnetic flux equivalent.

Inventors: TAJIMA KOUICHI, UMIDA HIDETOSHI, TAJIMA, KOUICHI, UMIDA, HIDETOSHI
Applicants: FUJI ELECTRIC CO LTD, FUJI ELECTRIC CO. LTD.
Application year: 1995
Main classification: H02P21
IPC codes: H02P27, H02P21