Motor system capable of obtaining high efficiency and method for controlling a motor

Abstract: In a brushless DC motor system, an RMS value of an input current of a voltage-fed inverter is detected by a current transformer and an RMS value detector circuit. A rotation speed of a brushless DC motor is detected based on a cycle of a position signal outputted by a position sensor circuit. Then, in response to the RMS value of the input current and the rotation speed, the phase of the inverter output voltage relative to the phase of the motor counter-electromotive voltage is set to a phase at which the motor efficiency comes generally to a peak, and a switching command to the voltage-fed inverter is generated. With this constitution, when the applied voltage waveform is controlled only by the voltage-fed inverter in response to the rotor-position sensor signal, the motor attains high-efficiency operation without involving any increase in cost and yet regardless of load conditions.

Description: TECHNICAL FIELD The present invention relates to a brushless DC, motor system capable of high efficiency operation as well as a control method for such a motor. BACKGROUND ART Brushless DC motors have conventionally been researched and developed for their application to various fields, and also put into practical use, taking advantage of their capability that no secondary copper loss is involved so that enhancement in efficiency is theoretically possible. The method for controlling brushless DC motors to high efficiency can be classified roughly into the following two: (i) A method in which the motor current is controlled so that the torque-to-current ratio is increased, by detecting the instantaneous value of the motor current; and (ii) A method in which a rotation speed of the motor is detected and the voltage-fed inverter is controlled in response to the detected rotation speed so that the inverter output voltage relative to the phase of the motor counter-electromotive voltage is brought to a phase that allows the brushless DC motor to be driven at around the maximum output. In the above (i) method, when a brushless DC motor in which a permanent magnet is mounted on the surface of the rotor (hereinafter, referred to as surface-magnet DC motor) is adopted, the motor current is controlled so that the d-axis current (a current in the same direction as that of the gap magnetic flux) that is irrelevant to torque generation is reduced to zero (see "Theory and Practical Design of AC servo systems," Sugimoto et al., published by Sogodenshi Shuppansha K.K., p. 74). Also, when a brushless DC motor in which a permanent magnet is mounted so as to be buried inside the rotor (hereinafter, referred to as buried-magnet DC motor) is adopted, the motor current is controlled based on the fact that the generated torque is the sum of magnet torque (a torque proportional to the q-axis current, which is a current in a direction perpendicular to that of gap magnetic flux) and reluctance torque (a torque proportional to the product of d-and q-axis currents), in such a way that optimum values for the d- and q-axis currents are obtained through the steps of detecting the load status (torque) and sequentially performing arithmetic operations to calculate the optimum values of the d- and q-axis currents (see "High Efficiency Control of Brushless DC Motors for Energy Saving," Morimoto et al., T.IEE Japan, Vol. 112-D, No. 3, '92, in particular, Equations (14) and (15)). The method (i) allows easy application to such fields as would demand high-speed torque response, for example, brushless DC motor drive systems used in machine tools, industrial robots, and the like, where a voltage-fed inverter is previously provided with an instantaneous current detector to implement a high-speed current control system so that there is no need of providing a special device for high efficiency. It is because the current response becomes generally equal to the torque response in brushless DC motors that the high-speed current control system is implemented. This allows a high-speed torque response to be realized. Meanwhile, the brushless DC motor drive system in household electrical appliances such as air conditioners, washers, and cleaners is based on a simple control system in which the voltage waveform is controlled by only a voltage-fed inverter in response to a position signal derived from detecting the position of the rotor of the brushless DC motor. Therefore, the above (i) method, if applied, would make it necessary not only to newly provide an instantaneous current detector but also to enhance the control operation function, particularly when a buried-magnet DC motor is adopted. As a result, a high-performance microcomputer needs to be substituted for or be added to an existing microcomputer, which incurs a great increase in cost. In view of these and other problems, it has been proposed to adopt such a constitution for brushless DC motor drive systems in household electrical appliances such as air conditioners, washers, and cleaners that the voltage-fed inverter is controlled so that the phase of the inverter output voltage relative to the phase of the motor counter-electromotive voltage will be set to a specified phase in response to a rotation speed of the brushless DC motor, in addition to the simple control system in which the voltage waveform is controlled by only a voltage-fed inverter in response to a position signal resulting from detecting the rotor position of the brushless DC motor. Adopting this constitution involves neither an instantaneous current detector nor a high-performance microcomputer, so that the aforementioned increase in cost can be largely reduced. As described above, with the constitution that the voltage-fed inverter is controlled so that the phase of the inverter output voltage relative to the phase of the motor counter-electromotive voltage will be set to a specified phase in response to a rotation speed of the brushless DC motor, the motor is enabled to attain high-efficiency operation as far as the operation is limited to around the rated point. However, the motor generally runs quite away from the rated point considerably longer than it does around the rated point. In the operation during the former period, the current value would not lower even if the voltage amplitude is reduced with a less load, so that a current more than necessary would be passed. As a result, the motor can be operated only at efficiencies lower than its best possible efficiency (peak efficiency at different load conditions) to a disadvantage. In more detail, if a buried-magnet DC motor with a d-axis inductance Ld of 65 mH and a q-axis inductance Lq of 15.0 mH and a counter-electromotive voltage coefficient Ke of 0.105 Vs/rad is driven at a rotation speed 90 rps., and if the phase of the motor counter-electromotive voltage and the phase of the inverter output voltage are set to 50° and 80°, respectively, then it can be understood from Fig. 1, which charts the motor output characteristic and motor current characteristic relative to the line voltage amplitude of the voltage-fed inverter, that although a brushless DC motor can be operated at larger output with larger settings of phase δ, the current value would show almost no decrease even if the line voltage amplitude of the voltage-fed inverter (amplitude of fundamental wave components) is reduced with lighter loads. Also, when the peak motor efficiency at different rotation speeds was measured with the load set to 10 kgf·cm and the phase δ adjusted by manual operation, a characteristic as marked by white circles in Fig. 2 was obtained. In contrast, when the motor efficiency with a low load at different rotation speeds was measured with a phase δ that is optimum for a high load (20 kgf·cm), a characteristic as marked by white triangles in Fig. 2 was obtained. Therefore, it can be understood that the motor efficiency lowers to a large extent when the motor is operated with a light load. Further, the larger the winding resistance is, the more noticeably the motor efficiency lowers. Fig. 3 is a block diagram schematically showing the arrangement of a conventional brushless DC motor system, and Fig. 4 is a block diagram showing the main part of the control circuit in Fig. 3. As shown in Fig. 3, the conventional brushless DC motor system is so arranged that an AC voltage 91 is converted into a DC voltage by a converter 92a of a voltage-fed inverter 92 and thereafter converted into an AC voltage by an inverter main part 92b and, as such, supplied to a brushless DC motor 93, where the magnetic-pole position of the rotor of the brushless DC motor 93 is detected by a position sensor circuit 94 and a switching command is generated by a control circuit 96 in response to a magnetic-pole position detection signal and then fed to the voltage-fed inverter 92. Also, as shown in Fig. 4, the control circuit 96 comprises an amplitude command output section 96a for receiving inputs of, for example, an actual rotation speed and a rotation speed command, which are obtained based on the time interval of magnetic-pole position detection signals outputted by the position sensor circuit 94, and a phase table 96b for outputting a phase command corresponding to the actual rotation speed by taking the actual rotation speed as a read address, whereby an amplitude command and a phase command are fed to an unshown PWM circuit so that a switching command is generated. The amplitude command output section 96a calculates, for example, a difference ΔY between a rotation speed command Y* and an actual rotation speed Y, and performs a PI operation of V* = Kp·ΔY + KI·σ(ΔY) by using the resulting difference ΔY, thereby determining and outputting an amplitude command V*, where Kp and,KI are each a constant and given by, for example, values obtained experientially. Therefore, a switching command is generated by feeding the amplitude command V* calculated from the difference between rotation speed command Y* and actual rotation speed Y as well as a phase command read from the phase table 96b to the PWM circuit, whereby the actual rotation speed Y can be approached and equalized to the rotation speed command Y*. However, when the above-described brushless DC motor control circuit is adopted, the motor efficiency with light loads would deteriorate to a large extent. Also, over the high-speed region in which the induced voltage of the brushless DC motor with no loads becomes larger than the inverter output voltage amplitude (a region in which the motor induced voltage is suppressed from increasing by utilizing the stator-to-armature reaction, that is, magnetic-flux weakening control is effected), the motor could not be controlled in such a way that the motor current is minimized by fixing the inverter voltage at a maximum. These points are now described in more detail. Fig. 5 shows an equivalent circuit of one phase portion of the brushless DC motor. As shown in the figure, this equivalent circuit has a motor winding resistance R, a reactance Xq on the q-axis, and an induced voltage (more precisely, a voltage resulting from adding the product of a difference between q-axis reactance Xq and d-axis reactance Xd and a d-axis component Id of motor current to the counter-electromotive voltage E) connected in series with the inverter fundamental-wave voltage V in this order. It is noted that j in Fig. 5 represents an imaginary number. In this arrangement, assuming that the frequency of inverter fundamental wave is ω (where the number of revolutions is ω /n in the case of a brushless DC motor with n pairs of magnetic poles), the inductances on the d- and q-axes are Ld and Lq, respectively, the counter-electromotive voltage coefficient is Ke, the motor current is I, the d-axis component and q-axis component of the motor current I are Id and Iq, respectively, the output torque is T, the d-axis component and q-axis component of the inverter fundamental-wave voltage V are Vd and Vq, respectively, and that the motor winding resistance R is substantially smaller than the magnitude |X| of the reactance, then the following relational equations are obtained: |I| = (Id 2 +Iq 2 ) 1/2 = [{(Vd/ω) /Xq} 2 + {(Vq/ω-Ke)/Xd} 2 ] 1/2 T = η{E+(Xq-Xd)·Id}·Iq/ω = η{Ke·Iq + (Lq-Ld)·Id·Iq} Then, on a brushless DC motor with motor constants Ld = 6.5 mH, Lq = 15.0 mH, Ke = 0.15 V·s/rad, when the output torque and the amplitude of motor current were calculated while the inverter fundamental-wave voltage amplitude |V| {= (Vd 2 +Vq 2 ) 1/2 } was varied, results as shown in Figs. 6A and 6B were obtained, where the number n of motor pole pairs was 2 and the rotation speed was set to 90 rps. As,apparent from Figs. 6A and 6B, such a reciprocal relationship is shown that the motor current amplitude increases even if the output torque T is decreased. That is, the joule loss of motor windings becomes greater with a light load (i.e., with a small output torque) than it is with a heavy load, so that the efficiency would lower considerably. Also, since the motor current amplitude becomes quite large with a light load, switching devices and the like with greater current capacities must be adopted as those constituting the voltage-fed inverter. Further, since the rotation speed of the brushless DC motor is controlled by adjusting the voltage amplitude of the voltage-fed inverter (PWM duty factor), the phase is fixed so that the voltage amplitude command will not be saturated at the time of acceleration in order to stably operate the speed control system. As a result, it is impossible to set such a phase that the voltage amplitude of the inverter is maximized at a constant speed. In other words, it is necessary to ensure margins for voltage amplitude adjustment at all times. Accordingly, it is impossible to perform such control that the motor current is minimized with the steady-state voltage amplitude of the voltage-fed inverter fixed at a maximum. The above-described disadvantages would be involved similarly in both cases of brushless DC motors in which a permanent magnet is disposed inside the rotor (buried-magnet motors), and brushless DC motors in which a permanent magnet is disposed on the surface of the rotor (surface-magnet motors). EP 0 427 571 A2 discloses a control device for a brushless motor comprising a source of DC power and inverter means for converting said DC power into AC power and supplying the AC voltage to the motor, further comprising a current detection means for detecting the current flowing from said DC power source to said inverter means and generating a signal representative of the current flow; power factor calculation means connected to the current detection means for calculating the power factor of the motor from the detected current and producing a corresponding power factor signal; voltage and frequency calculating means which receives as input signals the calculated power factor signal and a speed setting signal and produces control signals calculated from the input signals; and control means which receives the said control signals and generates drive signals to control the inverter means based on the control signals. Since a detection current signal of a current sampling and hold circuit is detected as current signals for each phase, a layout could therefore be adopted to perform for example PWM control of the inverter means, as conventionally, by obtaining a current command signal by detecting the difference between the speed command signal and the detected speed signal of the brushless motor. PWM control is then carried out in accordance with the magnitude of a deviation current value obtained by calculating the deviation between this current command signal which is thus obtained and currents flowing in each phase of the brushless motor. EP 0 363 073 A2 discloses a method and an apparatus for driving brushless motors. The method of driving an n-phase m-pole (n and m are integers) brushless motor comprises the steps of: detecting an harmonic component of back emf induced in the phases of the motor; deriving from the harmonic component phase energising signals; and using the phase energising signals to energise the phases of the motor in predetermined sequence. The step of detecting an harmonic component may include generating a simulated signal in response to the phase energising signals applied to the phases of the motor; providing from the motor and actual composite signal having an induced back emf component and a phase energising signal component; and obtaining the difference between the simulated and actual composite signals to produce a difference signal. The apparatus for driving an n-phase, m-pole (n and m are integers) brushless motor comprises: harmonic detecting means for detecting an harmonic component of back emf induced in the phases of the motor; phase energising means responsive to the harmonic component for generating phase energising signals to energise selected phases of the motor; and drive current means for supplying drive current to those phases energised by the phase energising signals. US 4 641 066 A discloses a control apparatus which detects a suitable starting point during rotational oscillation of a rotor at the start of a brushless motor by combining the polarity of a counter electromotive force generated in an armature coil of an armature winding of the brushless motor for a nonexcited phase due to the rotational oscillation of the rotor (when armature coils for particular phases are excited at the start of the brushless motor) with the pulse edge of an excitation switchover signal obtained from the potential of a neutral point of the armature winding having the armature coils connected in star-connection and that of a neutral point of a detecting resistance circuit having resistors connected in star-connection. Each of the resistors of the resistance circuit is arranged in parallel with a corresponding one of the armature coils the armature winding. By driving the brushless motor from the detected suitable starting point, it is possible to effect smooth and quick control of the starting operation of the brushless motor. DISCLOSURE OF THE INVENTION The present invention has been achieved in view of the aforementioned problems. A first object of the present invention is, therefore, in a brushless DC motor system in which the brushless DC motor is controlled by controlling the voltage waveform only with a voltage-fed inverter in response to a position signal, to attain high-efficiency operation regardless of load variations and to implement an extended range of operation and high-efficiency operation without additionally providing any special device, and yet to reduce the current capacity of switching devices or the like of the voltage-fed inverter. A further object of the present invention is to provide a DC motor system and an AC motor system which can be operated at peak efficiency while each motor can be prevented from stepping out. The present invention provides a method according to claim 1. In order to embody this control method, a brushless DC motor system according to claim 4 is provided which has an inverter control means for setting a switching command to the voltage-fed inverter in the way as described above. Preferred embodiments of the invention are indicated in the dependent claims. Therefore, according to the present invention, there is no need of margins for output voltage amplitude adjustment, such as in a case where the rotation speed is controlled by adjusting the output voltage amplitude of the voltage-fed inverter, so that the motor can be operated over a wider range by controlling the rotation speed with the output voltage amplitude of the voltage-fed inverter fixed to a maximum. Also, since the motor current amplitude can be reduced by reducing the output torque, the motor efficiency can be improved to a large extent at light loads. Further, since the motor current amplitude can be reduced, switching devices or the like of the voltage-fed inverter can be reduced in current capacity. Furthermore, since the conduction width of the voltage-fed inverter is 180° in electrical angle, the out-of-control period can be reduced to 0° in electrical angle. As a result, the motor terminal voltages can be increased, so that the motor can be operated over a wider range. Moreover, based on the fact that the motor terminal voltages can be increased, the amount of increase in motor current can be suppressed to small, so that the efficiency of the brushless DC motor can be enhanced by suppressing the increase in joule losses due to the motor windings, that the current can be passed in a desired direction over a range of 180° in electrical angle of the permanent magnet mounted on the rotor of the brushless DC motor, and further that the ratio of use of magnetic flux can be suppressed from lowering. Thus, the efficiency of the brushless DC motor can be enhanced. In one embodiment, the rotor of the brushless DC motor is a rotor having a permanent magnet disposed inside. In this case, not only a torque due to the magnet but also a torque due to the reluctance are generated, so that the generated torque on the whole can be increased without increasing the motor current. Also, the inductance of the motor windings can be increased (i.e., the magnetic-flux weakening effect can be increased) as compared with the surface-magnet motors, the motor can be operated with higher speeds than the surface-magnet motors. Further, based on the increased inductance of the motor windings, current ripples due to inverter lower-order harmonic-components can be reduced, so that torque ripples can also be reduced. According to one embodiment, a method for controlling a brushless DC motor comprises steps of obtaining a first neutral-point voltage by connecting resistors, which are connected at one end, respectively, to output terminals of individual phases of the voltage-fed inverter, to one another at the other end, obtaining a second neutral-point voltage by connecting stator windings of individual phases of the brushless DC motor to one another at one end, and detecting a magnetic-pole position of the rotor of the brushless DC motor based on a difference between the first neutral-point voltage and the second neutral-point voltage. A brushless DC motor system for embodying this method further comprises resistors which are connected to at one end, respectively, to output terminals of individual phases of the voltage-fed inverter and connected at the other end to one another, a difference-voltage output means for receiving, as inputs, a first neutral-point voltage obtained at the other ends of the resistors, and a second neutral-point voltage obtained at one end of stator windings of individual phases of the brushless DC motor at which the stator windings are connected to one another, and for delivering, as an output, a difference voltage between the two neutral-point voltages, and a rotor-position sensor means for detecting a magnetic-pole position of the rotor of the brushless DC motor based on the difference voltage. With this arrangement, the magnetic-pole position of the rotor can be detected regardless of rotation speed (except for at rest), conduction width, and current amplitude, and yet without providing any special sensor for detecting the magnetic-pole position of the rotor. When the brushless DC motor system according to any one of the aforementioned embodiments is adopted as a drive source in an electrical appliance, the motor can be operated over an expanded operational range without involving increase in size of the drive source due to the expansion of the operational range of the brushless DC motor as a drive source. Moreover, the power consumption can be reduced because of the enhanced efficiency of the brushless DC motor as a drive source. BRIEF DESCRIPTION OF THE DRAWINGS Fig. 1 is a chart showing the motor output and motor current characteristic versus the line voltage amplitude of the voltage-fed inverter; Fig. 2 is a chart showing results of measuring peak motor efficiencies at individual rotation speeds by manually adjusting the phase, and results of measuring the motor efficiency for low loads with individual rotation speeds at a phase optimum for high loads; Fig. 3 is a block diagram schematically showing the arrangement of a conventional brushless DC motor system; Fig. 4 is a block diagram showing part of the control circuit of the brushless DC motor system of Fig. 3; Fig. 5 is a diagram showing the equivalent circuit in a one-phase portion of a brushless DC motor; Figs. 6A and 6B are charts showing the torque-phase characteristic and the current amplitude-phase characteristic, respectively; Fig. 7 is a block diagram schematically showing a first embodiment of the brushless DC motor system of the present invention; Fig. 8 is a block diagram showing part of the control circuit of the brushless DC motor system of Fig. 7; Figs. 9A and 9B are charts representing the torque-phase characteristic and the current amplitude-phase characteristic, respectively; Fig. 10 is a chart showing running areas when the buried-magnet motor is driven by the control circuit of the first embodiment and driven by the conventional control circuit; Fig. 11 is a chart showing the efficiency-rotation speed characteristic when the buried-magnet motor is driven by the control circuit of the first embodiment and driven by the conventional control circuit; Fig. 12 is a block diagram schematically showing a second embodiment of the brushless DC motor system of the present invention; Fig. 13 is a block diagram showing the internal architecture of the microprocessor of Fig. 12; Fig. 14 is a flow chart for explaining in detail the processing contents of interrupt handling A shown in Fig. 13; Fig. 15 is a flow chart for explaining in detail the processing contents of interrupt handling B shown in Fig. 13; Fig. 16 is a diagram showing signal waveforms and processing contents of the individual components of the brushless DC motor system of Fig. 12; and Fig. 17 is a diagram showing signal waveforms of the individual components for explaining the position sensing operation by the amplifier, integrator, and zero-crossing comparator in the brushless DC motor system of Fig. 12. BEST MODE FOR CARRYING OUT THE INVENTION Embodiments of the present invention will now be described in detail with reference to the accompanying drawings. (First embodiment) Fig. 7 is a schematic block diagram of a brushless DC motor system of a first embodiment of the present invention. After an output voltage from an AC power supply 801 is converted into a DC voltage by a converter 802a in a voltage-fed inverter 802, the converted voltage is applied to an inverter main part 802b of the voltage-fed inverter 802, and an output voltage from the inverter main unit 802b is applied to a brushless DC motor (hereinafter, referred to simply as a motor) 803. The apparatus includes a position sensor circuit 804 which detects a rotor position of the motor 803 upon receiving an input of a voltage across terminals of the motor 803, and a control circuit 806 which generates a switching command upon receiving inputs of a position detection signal outputted from the position sensor circuit 804 and a rotation speed command value given externally and supplies the switching command to the inverter main unit 802b of the voltage-fed inverter 802. The control circuit 806 receives an input of a position detection signal outputted from the position sensor circuit 804 so as to calculate a phase of a motor counter-electromotive voltage, and calculates a cycle of the position detection signal so as to calculate a rotation speed of the motor 803 in response to the cycle. In response to the calculated cycle, rotation speed and rotation speed command value, a phase relative to the motor counter-electromotive voltage is set to a specified phase at which the motor 803 is operated at a peak efficiency under corresponding conditions, so that the phase can be achieved. In order to set a conduction period to 180° and achieve a specified output voltage amplitude determined uniquely with respect to the calculated rotation speed, the control circuit generates a switching command in response to the calculated rotation speed and outputs the switching command to the voltage-fed inverter 802. Fig. 8 is a block diagram of a part of the control circuit 806. The control circuit 806 includes an amplitude table 806a which receives an input (read address) of a rotation speed of the motor 803 and outputs a specified output voltage amplitude, and a phase command value calculator 806b which receives inputs of the rotation speed and the rotation speed command and performs a PI calculation of ΔY = Y*-Y and ϕ* = Kp·ΔY + KI·Σ (ΔY) so as to calculate a phase command ϕ* and output the resultant. It is to be noted that the output voltage amplitude and the phase command value are supplied to a PWM circuit (not shown). Further, in the above-mentioned equation, Y* is the rotation speed command, Y is a rotation speed, ΔY is a difference between the rotation speed command and the rotation speed, and Kp and KI are constants. Further, the rotation speed Y is an inverse number of a cycle X of the position detection signal. In the present case, look-up of the amplitude table 806a is effected every cycle in which the phase command is calculated by the phase command value calculator 806b. However, the look-up of the amplitude table 806a can be effected, for example, in accordance with a timing at which the rotation speed of the DC motor 803 varies. In the present embodiment, while the motor 803 is operated, a rotor position of the motor 803 is detected by the position sensor circuit 804 and then supplied to the control circuit 806. Then, a cycle of the position detection signal is calculated in the control circuit 806, and in response to the calculation of the cycle, the rotation speed is calculated. In the control circuit 806, a phase of voltage-fed inverter relative to the phase of the motor counter-electromotive voltage is set for the purpose of operating the motor at a peak efficiency under corresponding conditions, so that the phase is achieved. Also, a conduction period is set to 180°, and a switching command for achieving a specified output voltage amplitude determined uniquely with respect to the rotation speed is generated and supplied to the voltage-fed inverter 802. Consequently, the motor 803 can be operated in a state in which the phase of the output voltage of the voltage-fed inverter 802 relative to the motor counter-electromotive voltage is set to the above-mentioned phase and in which the output voltage amplitude is set to the specified output voltage amplitude determined uniquely with respect to the rotation speed, thereby allowing the motor efficiency to be the peak efficiency under the corresponding load conditions. Furthermore, the motor can be operated while fixing the output voltage amplitude of the inverter at maximum in a high-speed region in which the counter-electromotive voltage of the motor in the time of no load is greater than the output voltage amplitude of the voltage-fed inverter, thereby allowing the range of operation to be expanded. Also, a control for minimizing a motor current (increasing the efficiency) in the above-mentioned state can be performed. Fig. 9A is a graph of a torque to phase characteristic in a state in which an amplitude |V| of a fundamental wave of an output voltage of the voltage-fed inverter 802 that drives a buried-magnet motor of which d-axis inductance Ld and q-axis inductance Lq are respectively 6.5 mH and 15.0 mH, counter-electromotive voltage coefficient Ke is 0.15 V·s/rad and pole pair number n is 2 is set equal to a product of a rotation speed ω and the counter-electromotive voltage coefficient Ke. Fig. 9B is a graph of a current amplitude to phase characteristic in a state in which the output voltage fundamental wave amplitude |V| of the voltage-fed inverter 802 is set equal to the product of the rotation speed ω and the counter-electromotive voltage coefficient Ke. It can be understood that the torque and the current amplitude are increased by increasing the phase, and conversely the torque and the current amplitude are decreased by decreasing the phase. It is to be herein noted that the phase is given by tan -1 (Vd/Vq), while the torque T and the motor current amplitude are both given by the aforementioned equation. A case where the buried-magnet motor of which d-axis inductance Ld and q-axis inductance Lq are respectively 6.5 mH and 15.0 mH, counter-electromotive voltage coefficient Ke is 0.15 V·s/rad and pole pair number n is 2 is driven by the control apparatus of the present embodiment was compared with a case where the motor is driven by the prior art control apparatus in regard to their operation (i.e., running) areas, the comparison result being as shown in Fig. 10 (wherein a solid line indicates the case where the motor is driven by the control apparatus of the present embodiment, while a dotted line indicates the case where the motor is driven by the prior art control apparatus shown in Fig. 4). It can be understood that the operation area has been expanded to the high-speed side by driving the brushless DC motor by means of the control apparatus of the present embodiment. It is to be noted that, in the prior art control apparatus, a control system is designed so that the inverter maximum voltage in the steady-state stage (high-speed stage) is reduced by about 10% as compared with the control apparatus of the present embodiment taking a control margin into account. Further, Fig. 11 is a graph showing an efficiency to rotation speed characteristic in a case where a buried-magnet motor of which d-axis inductance Ld and q-axis inductance Lq are respectively 6.5 mH and 15.0 mH, counter-electromotive voltage coefficient Ke is 0.15 V·s/rad and pole pair number n is 2 is driven by the control apparatus of the present embodiment and in a case where the motor is driven by the prior art control apparatus shown in Fig. 4. In Fig. 11, a reference character A indicates a case where the motor is driven by the control apparatus of the present embodiment and a case where the motor is driven by the prior art control apparatus, a load torque being set to 20 kgf·cm in both cases (where the voltage amplitude is fixed in the control apparatus of the present embodiment, while the phase is fixed in the prior art control apparatus, so that an identical efficiency is achieved at the load torque of 20 kgf·cm). In Fig. 11, a reference character B indicates a case where the motor is driven by the control apparatus of the present embodiment when the load torque is set to 10 kgf·cm. In Fig. 11, a reference character C indicates a case where the motor is driven by the prior art control apparatus when the load torque is set to 10 kgf·cm. As apparent from Fig. 11, when the load torque is reduced, the efficiency is significantly reduced in the case where the motor is driven by the prior art control apparatus. However, in the case where the motor is driven by the control apparatus of the present embodiment, the necessary motor current is reduced together with the load torque, with which a Joule loss at the winding is reduced, resulting in slightly increasing the efficiency. Furthermore, in a magnetic-flux-weakened control area, a high-efficiency operation in which the output voltage amplitude of the voltage-fed inverter is maximized can be achieved, and the motor current is reduced by a degree corresponding to the increase of the efficiency. Therefore, a margin is produced in a current capacity of a switching element of the voltage-fed inverter 802 and so forth, consequently allowing the range of operation to be expanded. Conversely, when the range of operation is not expanded taking advantage of the margin of the current capacity, a switching element having a current capacity reduced by the margin can be employed, consequently achieving cost reduction. Also, no special device for performing the above-mentioned control is required to be incorporated, and therefore cost increase is prevented. (Second embodiment) Fig. 12 is a block diagram schematically showing a second embodiment of the brushless DC motor system of the present invention, while Fig. 13 is a diagram showing an internal structure of a microprocessor 918 shown in Fig. 12. In the present brushless DC motor system, three pairs of switching transistors 912u1, 912u2, 912v1, 912v2, 912w1 and 912w2 are serially connected in respective pairs to a DC power supply 911 (formed by serially connecting DC power supplies having mutually equal electromotive voltages and grounding a connection point of the sources) to form an inverter 912, and voltages at connection points of respective pairs of the switching transistors are applied respectively to star-connected stator windings 913u, 913v and 913w at respective phases of the brushless DC motor (hereinafter, referred to simply as a motor) 913. The voltages at the connection points of respective pairs of the switching transistors are also applied respectively to star-connected resistors 914u, 914v and 914w. Across the collector and emitter terminals of the switching transistors 912u1, 912u2, 912v1, 912v2, 912w1 and 912w2 are connected current feedback diodes 912u1d, 912u2d, 912v1d, 912v2d, 912wld and 912w2d. A reference numeral 913e denotes a rotor of the motor 913. Suffix characters u, v, and w correspond respectively to the u-phase, v-phase and w-phase of the motor 913. A voltage at a neutral point 913d of the star-connected stator windings 913u, 913v and 913w is supplied to an inverted input terminal of an amplifier 915 via a resistor 915a, while a voltage at a neutral point 914d of the star-connected resistors 914u, 914v and 914w are supplied to a non-inverted input terminal of the amplifier 915. Further, a resistor 915b is connected between an output terminal and the inverted input terminal of the amplifier 915, so that the amplifier 915 is made to operate as a differential amplifier. An output signal outputted from the output terminal of the amplifier 915 is supplied to an integrator 916 formed by serially connecting a resistor 916a and a capacitor 916b. An output signal from the integrator 916 (a voltage at a connection point of the resistor 916a and the capacitor 916b) is supplied to a non-inverted input terminal of a zero-crossing comparator 917 which receives at its inverted input terminal the voltage at the neutral point 913d. Therefore, a magnetic-pole position detection signal is outputted from an output terminal of the zero-crossing comparator 917. In other words, the aforementioned differential amplifier, integrator 916, and zero-crossing comparator 917 constitute a position sensor for detecting the magnetic-pole position of the rotor 913e of the motor 913. It is to be noted that a position sensor comprised of a rotary encoder and so forth can be adopted instead of the position sensor having the above-mentioned construction. The magnetic-pole position detection signal outputted from the position sensor is supplied to an external interrupt terminal of the microprocessor 918. In the microprocessor 918, an interrupt handling (interrupt handling A, see Fig. 14) is effected on a phase correction timer 918a and a cycle measuring timer 918b in response to the magnetic-pole position detection signal supplied to the external interrupt terminal. A timer value of the phase correction timer 918a is set by a timer value calculator 919a described later. The cycle measuring timer 918b supplies the timer value to a position signal cycle calculator 919b included in a CPU 919. The position signal cycle calculator 919b calculates a timer value per electrical angle of 1° based on a timer value corresponding to, for example, an electrical angle of 60°. The phase correction timer 918a supplies a count expiration signal to a 180° conduction inverter mode selector 919c included in the CPU 919 so as to execute an interrupt handling (interrupt handling B, see Fig. 15). The 180° conduction inverter mode selector 919c reads a corresponding voltage pattern from a memory 918c and outputs the same. In the CPU 919, a calculation based on the timer value is executed by the position signal cycle calculator 919b to output a position signal cycle signal to the timer value calculator 919a and a speed calculator 919e. The timer value calculator 919a calculates a timer value to be set in the phase correction timer 918a based on a position signal cycle signal from the position signal cycle calculator 919b and a differential speed from a phase command calculator 919f described hereinafter. The speed calculator 919e calculates the current-time speed based on the position signal cycle signal from the position signal cycle calculator 919b, and outputs a voltage command (output voltage amplitude command) based on the present speed while supplying the present speed to the phase command calculator 919f. The phase command calculator 919f, to which is also supplied a speed command, calculates a phase command by, for example, the aforementioned PI calculation based on the speed command and the present speed from the speed calculator 919e, and supplies the phase command to the timer value calculator 919a. Then, a voltage pattern outputted from the 180° conduction inverter mode selector 919c and a voltage command outputted from the speed calculator 919e are supplied to a PWM (Pulse Width Modulation) modulator 918d which outputs PWM modulation signals for the three phases. The PWM modulation signals are supplied to a base driver circuit 920, and the base driver circuit 920 outputs control signals to be supplied to the base terminals of the switching transistors 912u1, 912u2, 912v1, 912v2, 912w1 and 912w2. In the above description, as the components included in the CPU 919, only functional portions for achieving the corresponding functions are shown, and therefore the components are not existing inside the CPU 919 in a state in which the components can be clearly recognized. Here are shown voltage patterns corresponding to the inverter modes in Table 1. It is to be noted that the voltage patterns are represented by ON/OFF states of the switching transistors 912u1, 912u2, 912v1, 912v2, 912w1 and 912w2, where "1" corresponds to the ON-state, while "0" corresponds to the OFF-state. Inverter Voltage pattern Mode 912u1 912u2 912v1 912v2 912w1 912w2 0 1 0 0 1 1 0 1 1 0 0 1 0 1 2 1 0 1 0 0 1 3 0 1 1 0 0 1 4 0 1 1 0 1 0 5 0 1 0 1 1 0 Next, the operation of the brushless DC motor drive and control system of Fig. 12 will be described with reference to a waveform diagram shown in Fig. 16. In Fig. 16, as indicated in (A), (B) and (C), u-phase, v-phase and w-phase induced voltages Eu, Ev and Ew of the brushless DC motor vary as shifted in phase at intervals of 120°, and therefore a signal Vnm outputted from the amplifier 915 varies in a manner as shown in Fig. 16 (D), and an integral waveform fVnmdt of this signal integrated by the integrator 916 varies in a manner as shown in Fig. 16 (E). When the integral waveform is supplied to the zero-crossing comparator 917, an excitation switching signal that rises or falls at a zero-crossing point of the integral waveform is outputted in a manner as shown in Fig. 16 (F). The interrupt handling process A is executed with the rise and fall of the excitation switching signal, and the phase correction timer 918a starts (see start points (black dots) of arrows in Fig. 16 (G)). The timer value of the phase correction timer 918a is set by the time value calculator 919a, and the timer expires, or finishes counting after a time counting operation corresponding to a set timer value is completed (see end points of the arrows in Fig. 16 (G)). Then, every time the phase correction timer 918a expires, the interrupt handling process B is effected, and the 180° conduction inverter mode selector 919c increments the inverter mode by one step. That is, as shown in Fig. 16 (N), the inverter mode is selected in order of "1" "2" "3" "4" "5" "0" "1" "2" ···. Then, by incrementing the inverter mode by one step with the expiration of the phase correction timer 918a, the ON/OFF states of the switching transistors 912u1, 912u2, 912v1, 912v2, 912w1 and 912w2 are controlled in a manner as shown in Figs. 16 (H) through (M) in correspondence with each inverter mode. Consequently, the brushless DC motor 913 can be driven with the conduction period set at 180°, and the voltage-fed inverter voltage can be put in a state in which its phase precedes the phase of the motor induced voltage. In the present case, the amount of the precedence in phase of the voltage-fed inverter can be controlled by the phase correction timer 918a. Fig. 14 is a flow chart for explaining in detail the contents of the interrupt handling process A, where an external interrupt request is received at each of the leading edge and the trailing edge of the magnetic-pole position detection signal of the position sensor. Then, a value of the phase correction timer 918a is calculated at step SP1 based on a position signal cycle signal obtained by the position signal cycle calculator 919b and the differential speed obtained by the phase command calculator 919f, a correction timer value is set to the phase correction timer 918a at step SP2, and the phase correction timer is started at step SP3. Then, a cycle measuring timer started in the preceding interrupt handling process A is stopped at step SP4, and the cycle measuring timer value is read (stored) at step SP5. It is to be noted that the processes at step SP4 and step SP5 are processes for detecting the cycle of the edges of the magnetic-pole switching signal. Therefore, the cycle measuring timer is immediately reset for the next cycle measurement after reading a cycle measuring timer value, and then started. Then, a calculation of stored position signal cycle (e.g., calculation of the number of counts per electrical angle of 1°) is executed at step SP6, a current-time rotation speed of the motor 913 is calculated based on a result of the position signal cycle calculation at step SP7, a voltage command determined uniquely with respect to the current-time rotation speed is output at step SP8, and then the program flow returns to the original processing stage. Practically, when a count value corresponding to the interval of the magnetic-pole position detection signals is for example, 360 as a result of actual measurement by means of the cycle measuring timer 918a, a count value per cycle of the inverter output voltage is 360 × 6 = 2160 since the number of the inverter modes is six. Then, since the value of 2160 corresponds to 360°, a count value per 1° is 2160/360 = 6. When a phase amount command calculated based on the differential speed is 40°, the count value (timer value) corresponding to the phase amount command is 6 × (90-40) = 300. Therefore, the value of 300 is set in the phase correction timer 918a, and the phase correction timer 918a is started. Fig. 15 is a flow chart for explaining in detail the contents of the interrupt handling process B. When the phase correction timer 918a which has started during the interrupt handling process A expires, or finishes counting, the interrupt handling process B is effected. The inverter mode preparatorily set in the memory 918c is incremented by one step at step SP11, a voltage pattern corresponding to the incremented inverter mode is outputted at step SP12, and then the program flow returns to the original processing stage. A magnetic-pole position sensing operation executed by the amplifier 915, the integrator 916 and the zero-crossing comparator 917 shown in Fig. 12 will be described in more detail. In regard to the magnetic-pole position sensing effected by detecting the motor voltage, according to a magnetic-pole detection method using an induced voltage appearing in upper and lower arm-off intervals in a 120° conduction mode adopted in an electrical equipment such as an air conditioner, it is made impossible to detect an induced voltage when an electric current increases with a high load, and it is further made impossible to perform the magnetic-pole position sensing process. Assuming that a time to the cut-off of the current flowing through the stator winding is t, that an inverter electrical angle in the 180° interval is α[rad], and that an output frequency is f, a conditional expression for determining whether or not the detection of the induced voltage is enabled is: t < (π-α)/(4πf) As apparent from the above-mentioned conditions, it can be found that detection of the induced voltage is theoretically impossible when the 180° conduction is adopted. Further, when a greater torque is required, it is necessary to increase the current amplitude. However, when the current amplitude is increased, a great residual current due to a motor inductance results, and in a worst case, the current flowing through the stator winding in the 180° interval cannot be cut off. Therefore, in order to surely cut off the current flowing through the stator winding in the 180° interval, the current amplitude is required to be limited. Therefore, when the rotation speed is high or when the conduction time is prolonged, the current amplitude cannot be increased. However, when the construction shown in Fig. 12 is adopted, a voltage E N-0 at a neutral point 913d of the star-connected stator windings 913u, 913v and 913w is expressed by: E N-0 = (1/3){(V u-0 - E u-0 ) + (V v-0 - E v-0 ) + (V w-0 - E w-0 )} which represents a sum of 3n'th harmonic components (n: integer) (see Fig. 17 (G)) contained in the inverter output waveforms (see Fig. 17 (A), (B) and (C)) and the motor induced voltage waveforms (see Fig. 17 (D), (E) and (F)). Further, a voltage V M-0 at a neutral point 914d of the star-connected resistors 914u, 914v and 914w is expressed by: V M-0 = (1/3) (V u-0 + V v-0 + V w-0 ) (see Fig. 17 (H)). Therefore, by obtaining a difference E N-0 - V M-0 (see Fig. 17 (I)) of the voltages E N-0 and V M-0 , the 3n'th harmonic components contained in the motor induced voltage waveforms can be extracted. Furthermore, since each of the above-mentioned expressions does not depend on the current, there is no restriction on the conditional expressions, and therefore they can be applied to an arbitrary conduction period. That is, when the conduction period is set long at the high-speed rotation stage, the magnetic-pole position detection can be achieved without using a magnetic-pole position sensor even when the current amplitude is made great. In a case where the 180° conduction is adopted, also, the magnetic-pole position detection process can be achieved without using a magnetic-pole position sensor. The above has described only the brushless DC motor drive control in a state in which the brushless DC motor is running in the steady state. However, when the brushless DC motor is stopped, the aforementioned drive control cannot be effected because no induced voltage is generated then. Therefore, in starting the brushless DC motor operation, a voltage-fed inverter is compulsorily operated externally, and a three-phase AC voltage is applied, so that the rotor is rotated by a synchronous operation. Then, an induced voltage is generated when the rotor starts to rotate. Therefore the brushless DC motor can be subjected to the above drive control. Electrical appliances such as air conditioners, vacuum cleaners and electric washers are required to reduce the power consumption, and in recent years, brushless DC motors or inverters which have attracted public attention are being increasingly adopted. By applying the brushless DC motor drive system of the present invention to such electrical appliances, a power consumption is reduced further than in the conventional electrical appliances merely employing a brushless DC motor or an inverter. The above has described the case where a voltage-fed inverter is controlled by PWM. The present invention can also be applied similarly to a case where a voltage-fed inverter is controlled by PAM (a case where a switching element is provided in the converter 902a and the switching element is controlled to modulate a pulse amplitude). Furthermore, because a product of the rotation speed and the torque is the motor output, a torque control system can be constituted with incorporation of a torque sensor and so forth. INDUSTRIAL APPLICABILITY The brushless DC motors and other motors as well as their control methods according to the present invention are suited for use in household electrical appliances such as inverter air conditioners and other air conditioners, electric washers, and vacuum cleaners.

Claims: A method for controlling a brushless DC motor including steps of detecting a rotation speed and a rotor position of a brushless DC motor (803, 913) driven by a voltage-fed inverter (802, 912), characterized by the steps of: setting an output voltage amplitude of the voltage-fed inverter (802, 912) to a specified amplitude that is determined based on the detected rotation speed, and setting in response to a difference between a rotation speed command to the brushless DC motor (803, 913) and the detected rotation speed a switching command to the voltage-fed inverter (802, 912) such that the phase relative to the motor counter-electromotive voltage is set to a specified phase and such that a terminal voltage conduction width of the voltage-fed inverter (802, 912) is set to 180° in electrical angle. The method for controlling a brushless DC motor according to Claim 1, wherein the rotor of the brushless DC motor (913) is a rotor having a permanent magnet disposed inside. The method for controlling a brushless DC motor according to Claim 1, the method further comprising steps of obtaining a first neutral-point voltage by connecting resistors (914u, 914v, 914w), which are connected at one end, respectively, to output terminals of individual phases of the voltage-fed inverter (912), to one another at the other end, obtaining a second neutral-point voltage by connecting stator windings (913u, 913v, 913w) of individual phases of the brushless DC motor (913) to one another at one end, and detecting a magnetic-pole position of the rotor of the brushless DC motor (913) based on a difference between the first neutral-point voltage and the second neutral-point voltage. A brushless DC motor system in which a rotation speed and a rotor position of a brushless DC motor (803, 913) driven by a voltage-fed inverter (802, 912) are detected, and characterized by : an inverter control means (806, 918) for setting an output voltage amplitude of the voltage-fed inverter (802, 912) to a specified amplitude that is determined based on the detected rotation speed, and for setting in response to a difference between a rotation speed command to the brushless DC motor (803, 913) and the detected rotation speed a switching command to the voltage-fed inverter (802, 912) such that the phase relative to the motor counter-electromotive voltage is set to a specified phase and such that a terminal voltage conduction width of the voltage-fed inverter (802, 912) is set to 180° in electrical angle. The brushless DC motor system according to Claim 4, wherein the rotor of the brushless DC motor (913) is a rotor having a permanent magnet disposed inside. The brushless DC motor according to Claim 4, further comprising resistors (914u, 914v, 914w) which are connected at one end, respectively, to output terminals of individual phases of the voltage-fed inverter (912) and connected at the other end to one another; a difference-voltage output means (915) for receiving, as inputs, a first neutral-point voltage obtained at the other ends of the resistances (914u, 914v, 914w) and a second neutral-point voltage obtained at one end of stator windings (913u, 913v, 913w) of individual phases of the brushless DC motor (913) at which the stator windings are connected to one another, and for delivering, as an output, a difference voltage between the two neutral-point voltages; and a rotor-position sensor means (917) for detecting a magnetic-pole position of the rotor of the brushless DC motor (913) based on the difference voltage. An electrical appliance in which the brushless DC motor system according to Claim 4 is adopted as a drive source.

Inventors: KIMURA TAIZOU, KITANO NOBUKI, NISHIJIMA KIYOTAKA, OOYAMA KAZUNOBU, YAMAGIWA AKIO, YAMAI HIROYUKI, KIMURA, TAIZOU, KITANO, NOBUKI, NISHIJIMA, KIYOTAKA, OOYAMA, KAZUNOBU, YAMAGIWA, AKIO, YAMAI, HIROYUKI
Applicants: DAIKIN IND LTD, DAIKIN INDUSTRIES, LIMITED
Application year: 1995
Main classification: H02P6
IPC codes: H02P6