METHOD OF CONTROLLING DRIVING OF BRUSHLESS DC MOTOR, AND APPARATUS THEREFOR, AND ELECTRIC MACHINERY AND APPARATUS USED THEREFOR

Abstract: When a brushless D.C. motor 3, 13 is driven by voltage-fed inverters 2, 5, 12, 20, a conducting interval of the voltage-fed inverters 2, 5, 12, 20 is determined to be greater than 120° and equal or less than 180° by electrical angle so that improvement in efficiency and enlargement in operating range of the brushless D.C. motor is achieved with simple and cheap controlling.

Description: Technical Field The present invention relates to a brushless D.C. motor driving and controlling method and apparatus therefor and electric device. More particularly, the present invention relates to a brushless D.C. motor driving and controlling method and apparatus therefor which drive a brushless D.C. motor using voltage-fed inverters, and to an electrical equipment which employs a brushless D.C. motor as a driving source which is driven and controlled by the brushless D.C. motor driving and controlling apparatus. Background Art A brushless D.C. motor is known as a motor having higher efficiency in comparison to an A.C. motor. The brushless D.C. motor employs permanent magnets instead of rotor windings which magnets are equipped with a rotor. And, secondary copper losses caused by currents flowing in the rotor windings is made zero. It is known that a brushless D.C. motor has a remarkable efficiency improving effect in comparison to an A.C. motor when the brushless D.C. motor has a medium or small capacity which is equal or less than several tens kW. In a brushless D.C. motor driving system having such capacity, voltage-fed inverters or current control type inverters are employed as inverters for driving a brushless D.C. motor. Wherein, the current control type inverters has a main circuitry arrangement which is the same to that of voltage-fed inverters, and controls inverters so as to determine a motor current to be a desired value. A system which drives a brushless D.C. motor using the voltage-fed inverters, is mainly equipped with a device which is required improvement in saving the power consumption (improvement in efficiency), and is manufactured in large quantities, such as air conditioners, vacuum cleaners, electric washers and others. Therefore, a waveform of conducting interval which is 120° by electrical angle, is employed as a waveform control in inverters from the point of view of easiness in controlling. And, a system is employed which has a simple arrangement and which is cheap. (refer to "Microcomputer-Controlled Brushless Motor Without a Shift-Mounted Position Sensor", T.Endo et. al., IPEC-Tokyo'83,pp.1477-1488,1983: "Controlling Apparatus of a Brushless D.C. Motor", Nagata et. al., Japanese Patent Laid Open Gazett Tokuganhei 5-72197: "P.M.Brushless Motor Drives:A self-Commutating System without Rotor-Position Sensors", P.Ferraris et. al.,Ninth Annual Symposium-Incremental Motion Control Systems and Devices,pp.305-312,1980) Further, Magnetic pole position detection by detection of induced voltage of a motor, or a magnetic pole position detection sensor having a simple arrangement and employing Hall elements and the like is employed as an arrangement for detecting magnetic pole position of a brushless D.C. motor. A rotational position sensor which is expensive, such as a rotary encoder and the like, is not employed from the point of view of cost reduction. A system which drives a brushless D.C. motor using the current control type inverters, is applied to a machine tool, a servo-motor for an industrial robot and others which attach the most importance to the rapid torque response and the low torque ripple as required performances. In this system, a closed loop arrangement is employed in which a control circuitry determines (operates) output voltages of inverters so that current waveforms become desired waveforms, by giving attension to that currents and a torque of a brushless D.C. motor are functions of rotational position and are in proportional relationship. Therefore, the system has an expensive system arrangement which employs a precision sensor for detecting rotational position of a motor, current detectors for accurately controlling motor currents and a controller which can perform high-speed processing. Further, output voltages of inverters continuously vary because the inverters are controlled instantaneously in correspondence to a condition of a motor. In the system in which a brushless D.C. motor is driven using a conventional voltage-fed inverters, a switch of each phase of inverters is turned on for only an electrical angle of 120° in spite of positive and negative electrical angles are 180°, respectively. Therefore, the term of the rest electrical angle of 60° (the term of electrical angle of 120° for one cycle) is non-controlled condition. Consequently, inverters cannot output desired voltages during the term of non-controlled condition so that available rate of d.c. voltage in the inverters is low. And, terminal voltages of a brushless D.C. motor become small and an operating range becomes narrower (a maximum number of revolution becomes smaller) due to the low available rate of d.c. voltage. Further, motor currents should be increased for obtaining an output which is the same to that of a motor which has supplied not-lowered terminal voltages, because the terminal voltages are lowered. As a result, joule losses which are caused by resistances of motor windings, increase so that efficiency of a brushless D.C. motor cannot be improved to an expected degree. Although, permanent magnets are installed to a rotor of a brushless D.C. motor so that each permanent magnet corresponds to the electrical angle of 180°, currents can be flowed in desired directions for only an electrical angle of 120° so that available rate of magnetic flux is small. In other words, motor currents should be increased for obtaining a torque which is the same to that of a motor which is not lowered the available rate of magnetic flux. As a result, joule losses which are caused by resistances of motor windings, increase so that efficiency of a brushless D.C. motor cannot be improved to an expected degree. A method and an apparatus according to the preamble of claim 1 and claim 7 respectively is known. According to the known brushless D.C. motor has a fixed inner stator and a rotating outer rotor. The magnets for the motor are provided on the inner surface of the rotor. The US patent 4 641 066 discloses the arrangement of natural points and resistors in order to supply voltage to a brushless D.C. motor. The document IEEE Transactions on Industry applications vol. 24 (1988) Nov/Dec., N°6; B.k. BOSE : High- Performance Inverter-Fed Drive System of an Interior Permanent Magnet Synchronous Machine, pp 987 - 997 discloses an apparatus and an method according to Claims 1 an 7, wherein an conducting interval of 180° is achieved by clamping the value of the sinusoidal signal of the current to a predetermined value (quasi- square wave signal of Fig 12). Although excluded from the protection claimed, conducting interval of 180° are disclosed in the following. The deletion of these embodiments would be inappropriate for the clarity of the description. The present invention was made in view of the above-mentioned problems. It is an object of the present invention to supply a brushless D.C. motor driving and controlling method and apparatus therefor which improve available rates of voltage of voltage-fed inverters and motor flux using simple and cheap control, and which achieve higher efficiency and enlargement of operating range of a brushless D.C. motor. It is another object of the present invention to supply an electrical equipment which employs a brushless D.C. motor as a driving source which motor is driven and controlled by the brushless D.C. motor driving and controlling apparatus. These objects are solved by a brushless D.C. motor-driving and controlling method according to claim 1. Preferred developments of the method are given in the depending claims 2 to 6. The objects are also solved by a brushless D.C. motor-driving and controlling apparatus as given in claim 7. Preferred developments of the apparatus are given in the depending claims 8 to 13. As to the brush less D.C. motor driving and controlling method according to claim 1, the method determines a conducting interval of voltage-fed inverters to be a predetermined interval which is more than 120° or less than 180° by electrical angle when a brushless D.C. motor is driven by voltage-fed inverters. Therefore, the method determines non-control term to be less than 60° by electrical angle. As a result, the method increases motor terminal voltages and expands an operating range. Further, a quantity in increase of motor current is suppressed so that an increase of joule losses due to motor windings are suppressed and efficiency of the brushless D.C. motor is improved, because the motor terminal voltages can be increased. Further, a current can be forcibly flowed in a desired direction in correspondence to an extent of permanent magnets which is installed to the rotor of the brushless D.C. motor, the extent being greater than 120° by electrical angle. Therefore, lowering of available rate of magnetic flux is suppressed and efficiency of the brushless D.C. motor is improved. As to the brushless D.C. motor driving and controlling method according to claim 2, the method modulates outputs of voltage-fed inverters so as to output pulse signals, each pulse signal having constant pulse width within an e tire conducting interval. Therefore, magnetic pole position detection with high accuracy is not needed, and controlling is simplified. Also, the method improves efficiency and increases an amplitude of a fundamental wave in comparison to a case in which variable pulse width modulation is performed. The variable pulse width modulation varies pulse widthes of the pulse signal. Consequently, a maximum number of revolution of the brushless D.C. motor is increased. As to the method according again to claim 1, the method employs a rotor which includes permanent magnets in the interior of the rotor, as a rotor of a brushless D.C. motor. Therefore, the method generates not only a torque caused by magnet but also a torque caused by reluctance so that generated torque as a whole is increased without increasing motor currents. Further, inductance of motor windings is extremely increased in comparison to that of a brushless D.C. motor in which permanent magnets are installed on a surface of a rotor, so that operating with higher speed can be achieved in comparison to that of the brushless D.C. motor in which permanent magnets are installed on the surface of the rotor. Furthermore, the method decreases a current ripple due to low order higher harmonics components of inverters, because inductance of motor windings is great. Therefore, the method decreases a torque ripple. As to the method according to claim 3, the method controls voltage-fed inverters so that a phase of inverter output voltage is advanced from a phase of an induced voltage of a brushless D.C. motor, the latter phase being a phase which makes a brushless D.C. motor current and the induced voltage of the brushless D.C. motor to be the same phase to one another. Therefore, influence of inductance in a direction which is shifted by 90° electrically from a direction of magnetic flux generated by a permanent magnet, is efficiently and practically used. As a result, a current waveform is appoximated to be a sine wave. Consequently, torque ripple is further reduced and motor efficiency is further improved. Furthermore, reluctance torque and field weakening effect are efficiently and practically utilized. As to the method according to claim 4, the method obtains a first voltage at a neutral point by connecting one ends of resistors, the other end of each resistor is connected to each output terminal of voltage-fed inverters, obtains a second voltage at a neutral point by connecting one ends of stator windings of a brushless D.C. motor, and detects a magnetic pole position of a rotor of the brushless D.C. motor based upon a difference between the first voltage and the second voltage. Therefore, the magnetic pole position of the rotor is detected in spite of the revolution speed, conducting angle, and amplitude of motor current, without especially providing a sensor for detecting magnetic pole position of the rotor. As to the method according to claim 5, the method determines a conducting interval of voltage-fed inverters to be a predetermined interval which is more than 120° and less than 180° by electrical angle. Therefore, the difference signal between the voltages or an integrated signal which signals are used to detect magnetyic pole position of a motor rotor, is stabilized so that reliability is improved. As to the controlling method according to claim 6, the method determines a conducting interval of voltage-fed inverters to be a predetermined interval which is equal or more than 140° and equal or less than 170° by electrical angle. Therefore, motor efficiency and operating range are scarcely detracted. And, the difference signal between the voltages or an integrated signal is further stabilized so that reliability is further improved. As to the brushless D.C. motor driving and controlling apparatus according to claim 7, when a brushless D.C. motor is driven by voltage-fed inverters, the conducting interval determining means determins a conducting interval of voltage-fed inverters to be a predetermined interval which is more than 120° or less than 180° by electrical angle. Therefore, the apparatus determines non-control term to be less than 60° by electrical angle. As a result, the apparatus increases motor terminal voltages and expands an operating range. Further, a quantity in increase of motor currents is suppressed so that an increase of joule losses due to motor windings is suppressed and efficiency of the brushless D.C. motor is improved, because the motor terminal voltages can be increased. Further, a current can be forcibly flowed in a desired direction in correspondence to an extent of permanent magnets which is installed to the rotor of the brushless D.C. motor, the extent being greater than 120° by electrical angle. Therefore, lowering of available rate of magnetic flux is suppressed and efficiency of the brushless D.C. motor is improved. As to the controlling apparatus according to claim 8, the modulating means modulates outputs of voltage-fed inverters so as to output pulse signals, each pulse signal having constant pulse widthes within an entire conducting interval. Therefore, magnetic pole position detection with high accuracy is not needed, and controlling is simplified. Also, the apparatus improves efficiency and increases an amplitude of a fundamental wave in comparison to a case in which variable pulse width modulation is performed. The variable pulse width modulation varies pulse widthes of the pulse signal. Consequently, a maximum number of revolution of the brushless D.C. motor is increased. As to the apparatus according again to claim 7, the apparatus employs a rotor which includes permanent magnets in the interior of the rotor, as a rotor of a brushless D.C. motor. Therefore, the apparatus generates not only a torque caused by magnet but also a torque caused by reluctance so that generated torque as a whole is increased without increasing motor currents. Further, inductance of motor windings is extremely increased in comparison to that of a brushless D.C. motor in which permanent magnets are installed on a surface of a rotor, so that higher speed operating can be achieved in comparison to that of the brushless D.C. motor in which permanent magnets are installed on the surface of the rotor. Furthermore, the apparatus decreases a current ripple due to low order higher harmonics components of inverters, because inductance of motor windings is great. Therefore, the apparatus decreases a torque ripple. As to the apparatus according to claim 9, the phase controlling means controls voltage-fed inverters so that a phase of inverter output voltage is advanced from a phase of an induced voltage of a brushless D.C. motor, the latter phase being a phase which makes a brushless D.C. motor current and the induced voltage of the brushless D.C. motor to be the same phase to one another. Therefore, influence of inductance in a direction which is shifted by 90° electrically from a direction of magnetic flux generated by a permanent magnet, is efficiently and practically utilized. As a result, a current waveform is appoximated to be a sine wave. Consequently, torque ripple is further reduced and motor efficiency is further improved. Furthermore, reluctance torque and field weakening effect are efficiently and practically utilized. As to the apparatus according to claim 10, the apparatus obtains a first voltage at a neutral point by providing resistors which are connected their one ends to one another and are connected their other ends to each output terminal of voltage-fed inverters, and obtains a second voltage at a neutral point which is obtained at the connected ends of stator windings of a brushless D.C. motor which windings are connected to one another, and the difference voltage outputing means outputs a difference voltage between the first voltage and the second voltage, and the rotor position detecting means detects a magnetic pole position of a rotor of the brushless D.C. motor based upon the difference voltage. Therefore, the magnetic pole position of the rotor is detected in spite of the revolution speed, conducting angle, and amplitude of motor current, without especially providing a sensor for detecting magnetic pole position of the rotor. As to the apparatus according to claim 11, the apparatus employs a conducting interval determining means for determining a conducting interval of voltage-fed inverters to be a predetermined interval which is more than 120° and less than 180° by electrical angle. Therefore, the difference signal between the voltages or an integrated signal which signals are used to detect magnetic pole position of a motor rotor, is stabilized so that reliability is improved. As to the apparatus according to claim 12, the apparatus employs a conducting interval determining means for determining a conducting interval of voltage-fed inverters to be a predetermined interval which is equal or more than 140° and equal or less than 170° by electrical angle. Therefore, motor efficiency and operating range are scarcely detracted. And, the difference signal between the voltages or an integrated signal is further stabilized so that reliability is further improved. As to the electrical equipment according to claim 13, the equipment employs a brushless D.C. motor as a driving source which motor is driven and controlled by one of the brushless D.C. motor driving and controlling apparatus according to claims 9 through 16. Therefore, reducing in power consumption is performed due to the improvement of higher efficiency of a brushless D.C. motor which is the driving source. The present invention is described in more detail. When a brushless D.C. motor is driven by voltage-fed inverters which has a conducting interval of 120° by electrical angle, ideal current waveforms and an ideally generated torque are illustrated in Figs. 1(A) through 1(D). A brushless D.C. motor is a motor which replaces mechanical commutators of a d.c. motor with inverters. Therefore, when inverters are controlled so that each current of three phases (U, V, W) is joined at every 120° so as to make a motor current to be a direct current, a generated torque becomes similar to that of a d.c. motor, as is illustrated in Fig. 1(D). A simulation result is illustrated in Figs. 2(A) and 2(B) which simulation is carried out for a current waveform and generated torque when a brushless D.C. motor is driven by voltage-fed inverters which has a conducting interval of 180° by electrical angle. In this case, portions are generated in which a current of each phase overlaps to one another so that the brushless D.C. motor performs an operation which is different from that of a d.c. motor, because currents flows for a 180° term. As a result, ripple in the generated torque becomes great {refer to Fig. 2(B)}. From the above point of view, it is conventionally thought that voltage-fed inverters which has a conducting interval of 120° by electrical angle, is ideal for driving a brushless D.C. motor. Therefore, only a method which employs a conducting interval of 120° by electrical angle, is proposed and is available as a driving method of a brushless D.C. motor which driving method uses magnetic pole position detection based upon detection of induced voltages of a motor. But, when the inventors drives a brushless D.C. motor using voltage-fed inverters which has a conducting interval of 120° by electrical angle, the inventors have found that ideal waveforms illustrated in Figs. 1 are not obtained, and that waveforms illustrated in Figs. 3(A) and 3(B) are obtained, and that the waveforms illustrated in Figs. 3(A) and 3(B) are similar to the waveforms illustrated in Figs. 2(A) and 2(B). When Fig. 2(B) and Fig. 3(B) are compared to one another, it is understood that amplitudes of both torque ripple are in nearly the same degree to one another. It seems the reason that motor current cannot be controlled in a desired manner (controlled to be a rectangular waveform) when voltage-fed inverters employs a simple control is employed. Therefore, it becomes clear that lowering in efficiency and decreasing in operating range are not realized, the lowering and the decreasing being apprehended when a brushless D.C. motor is driven with a conducting interval of 180° by electrical angle. Rather, it becomes clear that efficiency of a brushless D.C. motor is improved by depression of increase in joule losses and depression of lowering in flux available rate. Further, when voltage-fed inverters which has a conducting interval exceeding 120° (other than 180°), portions are inevitably generated in which a current of each phase overlaps to one another so that a character seems to be obtained which character is similar to that of voltage-fed inverters with a conducting interval of 180° by electrical angle. Furthermore, in the voltage-fed inverters with a conducting interval of 120° by electrical angle, the character is improved by approximating each current waveform to a rectangular wave, as is illustrated in Figs. 1(A) through 1(C). On the contrary, in the voltage-fed inverters with a conducting interval of 180° by electrical angle, the character seems to be improved by smoothing each current waveform because each current flows in the entire interval. The present invention was made based upon the above findings. Brief Description of The Drawings Figures 1 are diagrams illustrating ideal current waveforms and generated torque under a condition that a brushless D.C. motor is driven by voltage-fed inverters with a conducting interval of 120° by electrical angle; Figures 2 are diagrams illustrating simulation results of a current waveform and generated torque under a condition that a brushless D.C. motor is driven by voltage-fed inverters with a conducting interval of 180° by electrical angle; Figures 3 are diagrams illustrating a current waveform and generated torque under a condition that a brushless D.C. motor is driven by voltage-fed inverters with a conducting interval of 120° by electrical angle; Figure 4 is a schematic block diagram illustrating an embodiment of a brushless D.C. motor driving and controlling apparatus according to the present invention; Figure 5 is a schematic cross sectional view of a brushless D.C. motor in which permanent magnets are installed on a surface of a rotor; Figure 6 is a schematic cross sectional view of a brushless D.C. motor in which permanent magnets are installed interior of a rotor; Figures 7 are graphs illustrating efficiency-revolution number characteristics and load torque-revolution number characteristics when a brushless D.C. motor having an arrangement illustrated in Fig. 5 is driven by voltage-fed inverters, and a conducting interval is determined to be 120°, 180° by electrical angle, respectively; Figures 8 are graphs illustrating efficiency-revolution number characteristics and load torque-revolution number characteristics when a brushless D.C. motor having an arrangement illustrated in Fig. 6 is driven by voltage-fed inverters, and a conducting interval is determined to be 120°, 150°, 180° by electrical angle, respectively; Figures 9 are graphs illustrating efficiency-revolution number characteristics and load torque-revolution number characteristics when a brushless D.C. motor having an arrangement illustrated in Fig. 6 is driven by voltage-fed inverters, and the inverter is modulated in a constant pulse width manner and variable pulse width manner, respectively by a control circuitry; Figures 10 are diagrams illustrating constant pulse width modulation and variable pulse width modulation; Figures 11 are diagrams illustrating a voltage waveform and current waveform when a brushless D.C. motor in which permanent magnets are installed on a surface of a rotor, is driven by voltage-fed inverters with a conducting interval of 180° by electrical angle and outputting constant pulse width modulation waveform; Figures 12 are diagrams illustrating a voltage waveform and current waveform when a brushless D.C. motor in which permanent magnets are installed interior of a rotor, is driven by voltage-fed inverters with a conducting interval of 180° by electrical angle and outputting constant pulse width modulation waveform; Figures 13 are diagrams illustrating a voltage waveform and a current waveform when a brushless D.C. motor in which permanent magnets are installed interior of a rotor, is supplied inverter output voltages which is advanced its phase from a phase which makes a phase of a brushless D.C. motor current and a phase of induced voltage of the brushless D.C. motor to be equal to one another; Figure 14 is a diagram schematically illustrating a brushless D.C. motor driving and controlling apparatus which achieves a conducting interval of 180° by electrical angle; Figure 15 is a diagram illustrating an inner arrangement of a microprocessor which is illustrated in Fig. 14; Figure 16 is a flowchart which explains processing contents of an interrupt handling 1 in detal; Figure 17 is a flowchart which explains processing contents of an interrupt handling 2 in detal; Figures 18 are diagrams illustrating signal waveforms and processing contents of each section of the brushless D.C. motor driving and controlling apparatus which is illustrated in Fig. 14; Figure 19 is a block diagram illustrating an inner arrangement of a microprocessor which is incorporated in a brushless D.C. motor driving and controlling apparatus which achieves a conducting interval of 150° by electrical angle; Figure 20 is a flowchart which explains processing contents of an interrupt handling 2' in detal; Figure 21 is a flowchart which explains processing contents of an interrupt handling 3 in detal; Figures 22 are diagrams illustrating signal waveforms and processing contents of each section of the brushless D.C. motor driving and controlling apparatus which is illustrated in Fig. 19; Figures 23 are diagrams illustrating signal waveforms of each section of the brushless D.C. motor driving and controlling apparatus which is illustrated in Fig. 14, which diagrams are useful in understanding position detection operation by an amplifier, integrator and zero-cross comparator; Figures 24 are diagrams illustrating a motor current and an integration signal of a position detector when a conducting interval of 180° by electrical angle, is employed; Figures 25 are diagrams illustrating a motor current and an integration signal of a position detector when a conducting interval of 120° by electrical angle, is employed; Figures 26 are diagrams illustrating a motor current and an integration signal of a position detector when a conducting interval of 130° by electrical angle, is employed; Figures 27 are diagrams illustrating a motor current and an integration signal of a position detector when a conducting interval of 140° by electrical angle, is employed; Figures 28 are diagrams illustrating a motor current and an integration signal of a position detector when a conducting interval of 150° by electrical angle, is employed; Figures 29 are diagrams illustrating a motor current and an integration signal of a position detector when a conducting interval of 160° by electrical angle, is employed; Figures 30 are diagrams illustrating a motor current and an integration signal of a position detector when a conducting interval of 170° by electrical angle, is employed; Figure 31 is a diagram illustrating operating ranges of a brushless D.C. motor when a conducting interval is determined to be 120°, 130°, 140°, 150°, 180° by electrical angle, respectively, and input currents are determined to be the same to one another; and Figure 32 is a diagram illustrating motor efficiencies of a brushless D.C. motor when a conducting interval is determined to be 120°, 130°, 140°, 150°, 180° by electrical angle, respectively, and input currents are determined to be the same to one another. Best Modes For Implementing The Invention Hereinafter, referring to the attached drawings, we explain the present invention in detal. Figure 4 is a schematic block diagram illustrating an embodiment of a brushless D.C. motor driving and controlling apparatus according to the present invention. Output voltages of inverters 2 are supplied to a brushless D.C. motor 3. A second voltage at a neutral point is obtained by Y-connecting induced voltages of the brushless D.C. motor 3 or by Y-connecting stator windings of each phase of the brushless D.C. motor 3. A first voltage at a neutral point is obtained by Y-connecting resistors between output terminals of each phase of the inverters 2. A motor position detection circuitry 4 receives a difference voltage between the first voltage and the second voltage. An output signal from the motor position detection circuitry 4 is supplied to a control circuitry 5. The control circuitry 5 generates a control command for determining a conducting interval to be more than 120° and equal to or less than 180° by electrical angle, and supplies the control command to the inverters 2. When the control circuitry 5 generates a control command for determining the conducting interval to be equal to 180° by electrical angle and supplies the control command to the inverters 2, The difference voltage should be supplied to the motor position detection circuitry 4, but when the control circuitry 5 generates a control command for determining the conducting interval to be less than 180° by electrical angle and supplies the control command to the inverters 2, the induced voltages or difference voltage may be supplied to the motor position detection circuitry 4. Therefore, a detection signal corresponding to a magnetic pole position of a motor rotor is obtained by the motor position detection circuitry 4 based upon the induced voltages of the brushless D.C. motor 3 or the difference voltage between the first and second voltages. The control circuitry 5 generates the control command based upon the magnetic pole position detection signal. The control command controls switches (not illustrated) of the inverters 2 so that the conducting interval is determined to be more than 120° by electrical angle or less than 180° by electrical angle. Figure 5 is a schematic cross sectional view of a brushless D.C. motor in which permanent magnets are installed on a surface of a rotor. Permanent magnets 3b are installed at predetermined position on a surface of a rotor 3a. A stator 3c includes a plurality of slots 3d in which stator windings (not illustrated) are wound. A d-axis which is indicated by an arrow in Fig. 5, is an axis indicating a direction of magnetic flux generated by the permanent magnet 3b. A q-axis is an axis which is shifted 90° electrically with respect to the d-axis. Figure 6 is a schematic cross sectional view of a brushless D.C. motor in which permanent magnets are installed interior of a rotor. Permanent magnets 3f are installed in a condition that the permanent magnets 3f are not exposed from a rotor 3e. Non-magnetic bodies 3g are installed between neighbouring permanent magnets 3f so that short-magnetic circuit between the neighbouring permanent magnets 3f is prevented from occurrence. An arrangement of a stator 3c is similar to that of the brushless D.C. motor illustrated in Fig. 5, therefore decription is omitted. Figures 7(A) and 7(B) are graphs illustrating efficiency-revolution number characteristics and load torque-revolution number characteristics when a brushless D.C. motor having an arrangement illustrated in Fig. 5 is driven by voltage-fed inverters, and a conducting interval is determined to be 120°, 180° by electrical angle, respectively. In Fig. 7(A), the characteristics are obtained under a condition that load torque is determined to be 20kg·cm. Also, in the figures, a represents characteristics when a conducting interval is determined to be 120° by electrical angle, and b represents characteristics when a conducting interval is determined to be 180° by electrical angle. As is apparent from these characteristics figures, efficiency is improved, a maximum revolution number is raised, and load torque in a high revolution range is increased, by determining the conducting interval to be 180° by electrical angle. Figures 8(A) and 8(B) are graphs illustrating efficiency-revolution number characteristics and load torque-revolution number characteristics when a brushless D.C. motor having an arrangement illustrated in Fig. 6 is driven by voltage-fed inverters, and a conducting interval is determined to be 120°, 150°, 180° by electrical angle, respectively. In Fig. 8(A), characteristics are obtained under a condition that load torque is determined to be 20kg·cm. Also, in the figures, a represents characteristics when a conducting interval is determined to be 120° by electrical angle, b represents characteristics when a conducting interval is determined to be 150° by electrical angle, and c represents characteristics when a conducting interval is determined to be 180° by electrical angle. As is apparent from these characteristics figures, efficiency is improved, a maximum revolution number is raised, and load torque in a high revolution range is increased, by determinng the conducting interval to be great. Figures 9(A) and 9(B) are graphs illustrating efficiency-revolution number characteristics and load torque-revolution number characteristics when a brushless D.C. motor having an arrangement illustrated in Fig. 6 is driven by voltage-fed inverters, and the inverters are modulated in a constant pulse width manner and variable pulse width manner, respectively by a control circuitry. In Fig. 9(A), characteristics are obtained under a condition that load torque is determined to be 20kg·cm and a conducting interval is determined to be 180° by electrical angle. Also, in the figures, a represents characteristics when variable pulse width modulation is performed, and b represents characteristics when constant pulse width modulation is performed. The variable pulse width modulation is a method for obtaining a voltage waveform which is equivalent to a sine wave and the like, by varying pulse widthes, as is illustrated in Fig. 10(A) . The constant pulse width modulation is a method which does not vary pulse widthes at all, as is illustrated in Fig. 10(B). As is apparent from these characteristics figures, efficiency is improved, a maximum revolution number is raised, and load torque in a high revolution range is increased, by employing the constant pulse width modulation. Hereinafter, description is made in more detail. A pulse width modulation method which employs variable pulse widthes such as sine wave modulation, is generally employed as other driving and controlling method of a motor and the like, because it is thought that low order higher harmonics waves cause increase in losses and vibration (refer to "Frequency Dependency of Induction Motor Parameters and Their Measuring Method", K.Kawagishi et. al., IPEC-Tokyo'83,pp.202-213,1983). When the method is applied to a brushless D.C. motor, a magnetic pole detection means with high accuracy is needed and control becomes complicated so that a cost as a whole is increased because a modulation signal waveform and a rotor should be synchronized to one another. But, the constant pulse width modulation does not have such disadvantages, and has an amplitude of a fundamental wave which is higher than that of a fundamental wave of variable pulse width modulation. Further, as is apparent from Figs. 9 which represent the inventors' knowledge, lowering in efficiency which is generally concerned, was not recognized, instead efficiency was improved. Furthermore, vibration was vibration which was almost inconsiderable when the method was taken in an air conditioner and the like. Next, a brushless D.C. motor in which permanent magnets are installed on a surface of a rotor, and a brushless D.C. motor in which permanent magnets are installed interior of a rotor, are described in detail (refer to Figs. 5 and 6). The brushless D.C. motor in which permanent magnets are installed on a surface of a rotor, which motor is illustrated in Fig. 5, includes permanent magnets 3b which are disposed on a surface of silicon steel plates of the rotor 3a. Therefore, an air gap (a distance between silicon steel plates of the rotor and silicon steel plates of a stator) is great so that inductance of stator windings become comparatively small. On the contrary, the brushless D.C. motor in which permanent magnets are installed interior of the rotor, which motor is illustrated in Fig. 6, includes permanent magnets 3b which are disposed interior of silicon steel plates of the rotor 3a. Therefore, an air gap is small so that inductance of stator windings become extremely great in comparison to that of the brushless D.C. motor in which permanent magnets are installed on the surface of the rotor, because the permanent magnets are installed interior of the rotor 3e. Figures 11 and 12 are diagrams illustrating a voltage waveform and current waveform when a brushless D.C. motor in which permanent magnets are installed on a surface of a rotor and a brushless D.C. motor in which permanent magnets are installed intrior of a rotor, is driven, respectively, by voltage-fed inverters with a conducting interval of 180° by electrical angle and outputting constant pulse width modulation waveform. In both figures, voltage waveforms are illustrated in upper portions, while current waveforms are illustrated in lower portions. As is apparent from both figures, due to effects of inductance of stator windings, a motor current becomes similar to a sine wave when a brushless D.C. motor in which permanent magnets are installed intrior of a rotor, is employed, so that losses caused by higher harmonic currents are reduced and efficiency is improved. Further, torque ripple is reduced. A brushless D.C. motor in which permanent magnets are installed on a surface of a rotor, and a brushless D.C. motor in which permanent magnets are installed intrior of a rotor, are further described in detail. The brushless D.C. motor in which permanent magnets are installed on a surface of a rotor, which motor is illustrated in Fig. 5, has cylindrical shaped silicon steel plates which constitute a rotor 3a. Therefore, inductance of stator windings are maintained to be constant regardless of rotational position of the rotor 3a. On the contrary, the brushless D.C. motor in which permanent magnets are installed intrior of a rotor 3e, which motor is illustrated in Fig. 6, has silicon steel plates which are magnetic bodies, and non-magnetic bodies 3g, which silicon steel plates and non-magnetic bodies 3g are alternately disposed in a portion of the rotor 3e which portion is close to an outer periphery of the rotor 3e. Therefore, inductance of stator windings vary depending upon rotational position of the rotor 3e. Specifically, when the brushless D.C. motor in which permanent magnets are installed on the surface of the rotor, and the brushless D.C. motor in which permanent magnets are installed intrior of the rotor, are operated in nominal conditions, both d-axis inductance and q-axis inductance of the former brushless D.C. motor are 3.2mH, while d-axis inductance and q-axis inductance of the latter brushless D.C. motor are 7.7mH, 22.8mH, respectively. A method for outputting inverter voltages in a matching-to-a-phase condition which phase maximize a d-axis voltage (induced voltage of the motor), that is for outputting inverter voltages so that a phase of a motor current and a phase of the induced voltage of the motor are the same to one another, is generally employed as a driving and controlling method of a brushless D.C. motor, because magnetic flux of permanent magnets are intended to be available at its maximum. In Figs. 11, a phase of inverter voltage is advanced by 29° with respect to a phase of induced voltage of a motor which motor is a brushless D.C. motor in which permanent magnets are installed on a surface of a rotor. Figs. 11 are obtained under a condition that a number of revolutions 2858 r.p.m., load torque is 20kg·cm, a motor voltage is 114.0V, and a motor current is 7.20A. In Figs. 11, scales are 200V/div. for voltage, 10A/div. for current, and 2msec/div. for horizontal axis. Further, the advanced angle varies depending upon motor constants. But, a brushless D.C. motor in which permanent magnets are installed interior of a rotor, has inductance of stator windings which is slightly greater than that of the former brushless D.C. motor. Therefore, a current waveform becomes a waveform which is illustrated in Figs. 12. Figs. 12 are obtained under a condition that a number of revolutions is 2858 r.p.m., load torque is 20kg·cm, a motor voltage is 126.5V, and a motor current is 5.76A. In Figs. 12, scales are 200V/div. for voltage, 10A/div. for current, and 2msec/div. for horizontal axis. A phase of inverter voltage is advanced by 70° with respect to a phase of an induced voltage of the motor in a case which is illustrated in Fig. 12. The advanced angle varies depending upon motor constants. When a phase of inverter voltage is further advanced with respect to the above phase of inverter voltage (advanced angle is 73°), and the inverter voltage is supplied to the brushless D.C. motor, a current waveform illustrated in lower portion of Figs. 13 is obtained. Figs. 13 are obtained under a condition that a number of revolutions is 2858 r.p.m., load torque is 20kg·cm, a motor voltage is 89.6V, and a motor current is 6.92A. In Figs. 13, scales are 200V/div. for voltage, 10A/div. for current, and 2msec/div. for horizontal axis. A voltage waveform is illustrated in upper portion of Figs. 13. When Figs. 12 and 13 are compared to one another, it is understood that the current waveform is quite similar to a sine wave due to effect of q-axis inductance. Therefore, torque ripple reducing effect and efficiency improving effect are further improved. Further, "reluctance torque" and "field weakening effect" which are other characteristics of a brushless D.C. motor in which permanent magnets are installed interior of a rotor, are effectively utilized. Figure 14 is a diagram schematically illustrating a brushless D.C. motor driving and controlling apparatus which achieves a conducting interval of 180° by electrical angle, while Figure 15 is a diagram illustrating an inner arrangement of a microprocessor which is illustrated in Fig. 14. Three pairs of switching transistors 12u1, 12u2, 12v1, 12v2, 12w1, 12w2 are serially connected, respectively, between terminals of a d.c. power source 11 so that inverters 12 are constituted. A voltage at each connecting point of each pair of switching transistors is supplied to each of three stator windings 13u, 13v, 13w of a brushless D.C. motor 13, the stator windings 13u, 13v, 13w being Y-connected and each of the stator windings 13u, 13v, 13w corresponding to each phase. Also, a voltage at each connecting point of each pair of switching transistors is supplied to each of three resistors 14u, 14v, 14w which are Y-connected. Further, diodes 12u1d, 12u2d, 12v1d, 12v2d, 12w1d, 12w2d for protection are connected between collector-emitter terminals of the switching transistors 12u1, 12u2, 12v1, 12v2, 12w1, 12w2, respectively. Furthermore, 13e represents a rotor of the brushless D.C. motor 13. Suffixes of u, v, w correspond to u-phase, v-phase, w-phase of the brushless D.C. motor 13, respectively. A second voltage of a neutral point 13d of the Y-connected stator windings 13u, 13v, 13w is supplied to a reversed input terminal of an amplifier 15 through a resistor 15a. A first voltage of a neutral point 14d of the Y-connected resistors 14u, 14v, 14w is supplied as it is to a non-reversed input terminal of the amplifier 15. A resistor 15b is connected between an output terminal and the reversed input terminal of the amplifier 15 so that the amplifier 15 operates as an differential amplifier. An output signal output from the output terminal of the amplifier 15 is supplied to an integrator 16 which is constituted by serially connecting a resistor 16a and a capacitor 16b. An output signal from the integrator 16 (a voltage at a connecting point of the resistor 16a and the capacitor 16b) is supplied to a non-reversed input terminal of a zero-cross comparator 17 which is supplied the second voltage of the neutral point 13d to its reversed input terminal. Therefore, a magnetic pole position detection signal is output from an output terminal of the zero-cross comparator 17. In other words, a position detector is constituted by the differential amplifier, integrator 16 and zero-cross comparator 17. A position detector which is constituted by a rotary encoder and the like may be employed instead of the position detector having the above arrangement. The magnetic pole position detection signal output from the position detector is supplied to an external interruption terminal of a microprocessor 18. In the microprocessor 18, an interrupt handling (refer to interrupt handling 1 in Fig. 15) for a phase correction timer 18a and a period measurement timer 18b, is carried out based upon the magnetic pole position detection signal which is supplied to the external interruption terminal. The phase correction timer 18a is determined its timer value by a timer value operating section 19a which is described later. The period measurement timer 18b supplies its timer value to a position signal period operating section 19b which is included in a CPU 19. The position signal period operating section 19b calculates a timer value per 1 cycle by electrical angle based upon a timer value corresponding to 60° by electrical angle, for example. The phase correction timer 18a supplies a count over signal to inverter mode selection section 19c for a conducting interval of 180° by electrical angle so that an interrupt handling (refer to an interrupt handling 2 in Fig. 2) is carried out. The inverter mode selection section 19c for a conducting interval of 180° by electrical angle, reads out corresponding voltage pattern from a memory 18c and outputs therefrom. In the CPU 19, the position signal period operating section 19b performs operation based upon the timer value and outputs a position signal period signal which is supplied to a timer value operating section 19a and a velocity operating section 19e. The timer value operating section 19a is also supplied a phase angle command. The timer value operating section 19a calculates a timer value which is determined in the phase correction timer 18a, based upon the phase angle command and the position signal period signal from the position'signal period operating section 19b. The velocity operating section 19e calculates a velocity at the present time based upon the position signal period signal from the position signal period operating section 19b, and supplies the calculated velocity to a velocity controlling section 19f. The velocity controlling section 19f is also supplied a velocity command. The velocity controlling section 19f outputs a voltage command based upon the velocity command and the velocity at the present time from the velocity operating section 19e. And, the voltage pattern output from the inverter mode selection section 19c for a conducting interval of 180° by electrical angle, and the voltage command output from the velocity controlling section 19f are supplied to a PWM (Pulse Width Modulation) modulation section 18d. The PWM modulation section 18d outputs PWM modulation signals for three phases. The PWM modulation signals are supplied to a base driving circuitry 20. The base driving circuitry 20 outputs control signals for supplying base terminals of the switching transistors 12u1, 12u2, 12v1, 12v2, 12w1, 12w2. Each section included in the CPU 19 represents a functional portion for performing corresponding function. These sections do not exist in the CPU 19 in a recognizable condition. The Voltage patterns corresponding to inverter modes are represented in table 1. The voltage patterns are represented with ON-OFF conditions of the switching transistors 12u1, 12u2, 12v1, 12v2, 12w1, 12w2. "1" corresponds to ON condition, while "0" corresponds to OFF condition. Inverter mode Voltage-Fed inverters 12u1 12u2 12v1 12v2 12w1 12w2 0 1 0 0 1 1 0 1 1 0 0 1 0 1 2 1 0 1 0 0 1 3 0 1 1 0 0 1 4 0 1 1 0 1 0 5 0 1 0 1 1 0 Next, referring to waveform diagrams illustrated in Figs. 18, operation of the brushless D.C. motor driving and controlling apparatus which is illustrated in Fig. 14, is described. The u-phase, v-phase and w-phase induced voltages Eu, Ev, Ew of the brushless D.C. motor vary under a condition that three induced voltages are sequentially shifted their phase by 120°, as are illustrated in Figs. 18(A), 18(B) and 18(C). The signal Vnm output from the amplifier 15 varies, as is illustrated in Fig. 18(D). An integration waveform {formula (1)} which is obtained by performing integration of the signal Vnm by the integrator 16, varies , as is illustrated in Fig. 18(E). ∫ V n m d t The integration waveform is supplied to the zero-cross comparator 17, and the zero-cross comparator 17 outputs a exitation change-over signal which rises or falls at zero-cross points of the integration signal, as is illustrated in Fig. 18(F). The interrupt handling 1 is performed due to the rising and falling of the excitation change-over signal so that the phase correction timer 18a starts {refer to start points of arrows illustrated in Fig. 18(G)}. The phase correction timer 18a performs time counting operation for the determined timer value and count-over is generated {refer to end points of arrows illustrated in Fig. 18(G)}, because the phase correction timer 18a is determined its timer value by the timer value operating section 19b. The interrupt handling 2 is carried out at every generation of count-over of the phase correction timer 18a, and the inverter mode selection section 19c for a conducting interval of 180° by electrical angle, advances the inverter mode by 1 step. That is, the inverter mode is selected in the order of "1", "2", "3", "4", "5", "0", "1", "2", ..., as is illustrated in Fig. 18(N). The inverter mode is advanced by 1 step due to the count-over of the phase correction timer 18a, so that the switching transistors 12u1, 12u2, 12v1, 12v2, 12w1, 12w2 are controlled their ON-OFF conditions in correspondence to each inverter mode, as are illustrated in Figs. 18(H) through 18(M). As a result, driving of the brushless D.C. motor 13 is performed under a condition that the conducting term is determined to be 180° by electrical angle, and the phase of the inverter voltage is determined to be an advanced phase with respect to the induced voltage of the motor. Wherein, an advancing angle of the phase of the inverter voltage is controlled by the phase correction timer 18a. Figure 16 is a flowchart which explains the contents of processing of the interrupt handling 1 in detail. An external interruption request is accepted at a rising edge and falling edge, respectively, of a magnetic pole position detection signal (which signal corresponds to the exitation change-over signal) of the position detector. In step SP1, a timer value of-the phase correction timer 18a is calculated based upon an externally supplied phase angle (phase correction angle) command and the position signal period signal obtained by the position signal period operating section 19b. In step SP2, the phase correction timer 18a is set with a correction timer value (timer value for correction). In step SP3, the phase correction timer 18a is started. In step SP4, the period measurement timer 18b is stopped which was started in the previous interrupt handling 1. In step SP5, The timer value of the period measurement timer 18b is read into (is stored). These,processings in steps SP4 and SP5 are processings for detecting a period of an edge of the exitation change-over signal. Therefore, the period measurement timer 18b is reset and started immediately after the reading of the timer value of the period measurement timer 18b, for the next period measurement. In step SP6, an operation of the stored position signal period (for example, calculation of a counting number per 1° by electrical angle). In step SP7, the revolution number at the present time of the brushless D.C. motor 13 is calculated based upon the position signal period operation result. In step SP8, velocity controlling is performed so as to follow the externally supplied velocity command, and the voltage command is output. Thereafter, processing is returned to prior processing. Specifically, when the count value corresponding to the interval of the magnetic pole position detection signal, is 360 which is obtained by the actual measurement by the period measurement timer 18b, the count value per 1 period of the inverter output voltage becomes 360 x 6 = 2160 because a number of inverter modes is 6. And, the count value for 1° becomes 2160/360 =6 because the count value of 2160 corresponds to 360°. When the phase angle command is 60°, the count value (timer value) corresponding to the phase angle command becomes 6 x 60 =360. Therefore, the value of 360 is set in the phase correction timer 18a as the timer value for correction, and the phase correction timer 18a is started. Figure 17 is a flowchart which explains contents processing of the interrupt handling 2 in detail. The interrupt handling 2 is accepted when the phase correction timer 18a becomes count-over condition which timer was started in the interrupt handling 1. In step SP1, the inverter mode is advanced by 1 step which inverter mode is previously determined in the memory 18c. In step SP2, the voltage pattern corresponding to the advanced inverter mode is output. Thereafter, processing is returned to prior processing. Therefore, a number of timers is decreased by 1 and interrupt handlings are decreased by 1 when the above controlling is realized using a microcomputer, which advantages will be apparent by comparing the above embodiment to a comparative example which is described below. Further, when the above contolling is realized using a hardware, a number of counters is decreased by 1. Comparative Example Figure 19 is a diagram illustrating an inner arrangement of the microprocessor 18 which is incorporated in a brushless D.C. motor driving and controlling apparatus for realizing a conducting interval of 150° by electrical angle. The microprocessor 18 realizes the conducting interval of 150° by electrical angle. The inner arrangement differs from the inner arrangement which is illustrated in Fig. 15 in that an inverter mode selection section 19c' for a conducting interval of 150° by electrical angle, is employed instead of the inverter mode selection section 19c for a conducting interval of 180° by electrical angle, a second phase correction timer 18e is further provided, and contents of the interrupt handling 2' and a number of inverter modes are increased. The second phase correction timer 18e is started in the interrupt handling 2' which is caused by the phase correction timer 18a', and supplies its count-over signal to the inverter mode selection section 19c' for a conducting interval of 150° by electrical angle which section is included in the CPU 19' so that an interrupt handling (refer to the interrupt handling 3 in Fig. 19) is carried out. In this comparative example, component sections corresponding to the component sections illustrated in Fig. 15, are applied reference codes which are added "'". Further, arrangements and interconnections of the inverter circuitry, brushless D.C. motor, Y-connected resistors, position detector, base driving circuitry are the same to the arrangements and interconnections illustrated in Fig. 14, therefore illustration and description are omitted. The second phase correction timer 18e is determined its timer value by the timer value operating section 19a'. Voltage patterns corresponding to inverter modes are represented in table 2. The voltage patterns are represented with ON-OFF conditions of the switching transistors 12u1, 12u2, 12v1, 12v2, 12w1, 12w2. And, "1" corresponds to ON condition, while "0" corresponds to OFF condition. Inverter mode Voltage-Fed inverters 12u1 12u2 12v1 12v2 12w1 12w2 0 1 0 0 1 1 0 1 1 0 0 1 0 0 2 1 0 0 1 0 1 3 1 0 0 0 0 1 4 1 0 1 0 0 1 5 0 0 1 0 0 1 6 0 1 1 0 0 1 7 0 1 1 0 0 0 8 0 1 1 0 1 0 9 0 1 0 0 1 0 10 0 1 0 1 1 0 11 0 0 0 1 1 0 Figs. 22 are waveform diagrams of component sections which waveforms represent the operation of this comparative example. Waveforms illustrated in Figs. 22(A) through 22(G) are the same to those of Figs. 18(A) through 18(G). And, voltage patterns corresponding to inverter modes which correspond to even numbers, are output b'y the interrupt handling 2' and voltage patterns corresponding to inverter modes which correspond to odd numbers, are output by the interrupt handling 3, due to the addition of Fig. 22(G'), so that driving of a brushless D.C. motor under a condition that the conducting term is determined to be 150° by electrical angle, is performed. Fig. 20 is a flowchart which explains contents of processing of the interrupt handling 2' in detail. The interrupt handling 2' is accepted when the phase correction timer 18a' becomes count-over condition which timer is started in the interrupt handling 1'. In step SP1, an inverter mode is advanced by 1 step, which inverter mode is previously determined in the memory 18c'. In step SP2, a voltage pattern corresponding to the advanced inverter mode is output. In step SP3, a timer value (a value corresponding to 30°) for the second phase correction timer 18e is calculated based upon the phase correction value. In step SP4, the timer value for correction is set to the second phase correction timer 18e. In step SP5, the second phase correction timer 18e is started. Thereafter, processing is returned to prior processing. Fig. 21 is a flowchart which explains contents of processing of the interrupt handling 3 in detail. The interrupt handling 3 is accepted when the second phase correction timer 18e becomes count-over condition, which timer is started in the interrupt handling 2'. In step SP1, an inverter mode is advanced by 1 step, which inverter mode is previously determined in the memory 18c'. In step SP2, a voltage pattern corresponding to the advanced inverter mode is output. Thereafter, processing is returned to prior processing. The magnetic pole position detection by the amplifier 15, integrator 16 and zero-cross comparator 17 which are illustrated in Fig. 14, is described in more detail. In the magnetic pole position detection which is achieved by detecting motor voltages, a magnetic pole position detection method is employed in an electric equipment such as an air conditioner or the like, which method uses induced voltages which appear in both upper and lower arms' OFF term of the conducting interval of 120° by electrical angle. The method becomes impossible to detect the induced voltages when high load is employed and currents are increased so that the magnetic pole position detection becomes impossible to be achieved. When it is assumed that time period until the flowing current flowing in stator windings being cut off, is t, conducting angle of inverters in the term of 180° by electrical angle is determined to be α (rad), and output frequency is f, a condition formula for determining possibility and impossibility of induced voltage detection, is as follows. t < (π -α) / (4πf) As is apparent from the condition formula, it is understood that detection of induced voltage is principally impossible when a conducting interval of 180° by electrical angle is employed. When a greater torque is required, an amplitude of a current should be increased. When an amplitude of a current is increased, a residual current due to motor inductance becomes greater. In the worst case, a current flowing in a stator winding is not cut off during a 180° term (a term corresponding to 180° by electrical angle). Therefore, an amplitude of a current should be limited so that a current flowing in a stator winding is securely cut off during the 180° term. Consequently, an amplitude of a current cannot be increased too much when high speed revolution is performed and/or when conducting period is lengthened. When the arrangement illustrated in Fig. 14 is employed, the voltage E N-0 at the neutral point 13d of the Y-connected stator windings 13u, 13v, 13w becomes the following equation: E N-O =(1/3){(V u-0 - E u-0 ) + (V v-0 - E v-0 ) + (V w-0 - E w-0 )} In other words, the voltage E N-0 becomes a sum {refer to Fig. 23(G)} of 3n order higher harmonic components (wherein, n is an integer) which are included in the inverter output waveforms {refer to Figs. 23(A), 23(B) and 23(C)} and the induced voltage waveforms {refer to Figs. 23(D), 23(E) and 23(F)} of a motor. Further, the voltage V M-0 at the neutral point 14d of the Y-connected resistors 14u, 14v, 14w becomes the following equation {refer to Fig. 23(H)}: V M-0 = (1/3) (V u-0 + V v-0 + V w-0 ) Therefore, the 3n order higher harmonic component is taken out by obtaining a difference E N-0 -V M-0 {refer to Fig. 23(I)} between the both voltages E N- 0 , V M-0 . The arrangement illustrated in Fig. 14 is not limited by the condition formula and is applicable to conducting term having an arbitrary angle, because the both equations are independent of a current. That is, magnetic pole position detection is achieved without using a magnetic pole position sensor especially when high speed revolution is performed and/or when conducting period is lengthened and when an amplitude of a current is increased. Also, magnetic pole position detection is achieved without using a magnetic pole position sensor especially when a conducting interval of 180° by electrical angle is employed. In the foregoing, driving and controlling of a brushless D.C. motor is described under only a condition that the brushless D.C. motor is under steady operation. But, the above driving and controlling cannot be achieved when the brushless D.C. motor is stopped, because no induced voltages are generated. Therefore, when driving of the brushless D.C. motor is started, three-phase alternating voltage is externally and forcibly supplied to the brushless D.C. motor so that the rotor is rotated by synchronized operation. When the rotor begins its revolution, an induced volatge is generated so that the above driving and controlling of the brushless D.C. motor is achieved. Further, when the conducting interval of 180° by electrical angle is employed, and when a brushless D.C. motor in which permanent magnets are installed interior of a rotor which motor has a special application, is driven under a condition of high revolution and great load, the motor current and the integration signal of the position detector are varied as are illustrated in Figs. 24(A) and 24(B). It is understood that the integration signal is greatly disorder when the conducting interval of 180° by electrical angle is employed, by comparing the motor current and the integration signal to the motor current and the integration signal {refer to Figs. 25(A) and 25(B)} which are obtained when the conducting interval of 120° by electrical angle is employed. Of course, a difference signal between voltages is disorder. Therefore, when a slight varying in load, resonance and the like in the driving system of the inverters and the brushless D.C. motor are generated, a point is appeared at which the integration signal is not zero-crossed due to the disorder so that the position signal is bot obtained and the brushless D.C. motor may step out. The motor current illustrated in Fig. 24(A) is different from the current waveforms illustrated in in Figs. 2(A), 12(B) and 13(B). The difference is caused by difference in motor application. Provided that such disadvantage are rarely generated. To dissolve such disadvantage which is generated under a special condition, so that reliability of the apparatus for driving and controlling the brushless D.C. motor is improved, the conducting interval should be determined to be more than 120° and less than 180° by electrical angle. Figs. 26 through 30 illustrate motor currents and integration signals which correspond to cases that the conducting interval is determined to be 130°, 140°, 150°, 160°, and 170° by electrical angle, respectively. As is apparent from those figures, disorder in integration signals are eliminated. Further, operating ranges of a compressor ( a compressor which employs a brushless D.C. motor as a driving source), and motor efficiencies of the brushless D.C. motor are illustrated in Figs. 31 and 32 which correspond to cases that the conducting interval is determined to be 130°, 140°, 150°, and 180° by electrical angle, respectively, and that the input currents are determined to be the same to one another. As is apparent from Fig. 31, the operating ranges scarcely vary when the conducting interval is determined to be 140°-170° by electrical angle. Also, as is apparent from Fig. 32, lowering in the motor efficiency is less than 1% when the conducting interval is determined to be 140°-170° by electrical angle. Therefore, it is preferable that the conducting interval is determined to be 140°-170° by electrical angle. In this case, the above disadvantage is securely prevented from occurrence with scarcely reducing of the operating range and with scarcely lowering the motor efficiency, so that the reliability is remarkably improved. Further, an arrangement may be employed instead of the position detection circuitry which includes the amplifier 15, integrator 16 and zero-cross comparator 17. The arrangement is illustrated in Fig. 33(A), and has an amplifier 115 which is supplied the voltage at the neutral point 13d (refer to Fig. 14) to a non-reversed input terminal through a resistor 115a, is supplied the voltage at the neutral point 14d (refer to Fig. 14) to a reversed input terminal. And, a resistor 115b and a capacitor 116b are connected in parallel between an output terminal and the reversed input terminal of the amplifier 115. Furthermore, an arrangement which is illustrated in Fig. 33(B) may be employed. The arrangement is obtained by adding an amplifier 117 to the arrangement illustrated in Fig. 33(A), which amplifier 117 has a resistor which is connected between an output terminal and a reversed input terminal. In this arrangement, an output signal from the amplifier 115 is supplied to the reversed input terminal of the amplifier 117 through a resistor. Further, an arrangement which is illustrated in Fig. 33(C) may be employed. The arrangement is obtained by replacing the amplifiers 115 and 117 to one another, which amplifiers appear in Fig. 33(B). When the arrangement illustrated in Fig. 33(C) is employed, the voltage at the neutral point 13d may be supplied to the reversed input terminal of the amplifier 117 through a resistor and the voltage at a neutral point 14d may directly be supplied to a non-reversed input terminal of the amplifier 117. When one of those arrangements is employed, operation which is similar to that of a corresponding portion which is illustrated in Fig. 14 is achieved. Further, reducing in power consumption is required for an electrical equipment such as air conditioner, vacuum cleaner, electrical washer and the like, and a brushless D.C. motor and an inverter which are drawn attention in recent years, are going to be employed. When the apparatus for driving and controlling a brushless D.C. motor according to the present invention is applied to those electrical equipments, power consumption is further reduced in comparison to conventional electrical equipmenats which employ a brushless D.C. motor and inverters. Possibility in Industrial Utilization The present invention performs improvement in efficiency and enlargement in operating range of a brushless D.C. motor which is employed as a driving source in various applications.

Claims: A brushless D.C. motor driving and controlling method which drives a brushless D.C. motor (3)(13) using voltage-fed inverters (2)(5)(12)(20), wherein a conducting interval of voltage-fed inverters is determined to be a predetermined interval which is more than 120° and less than 180° by electrical angle, characterized in that a rotor (3e) is employed which includes permanent magnets (3f) in the interior of said rotor (3e), as rotor (3e) of said brushless D.C. motor (3) (13). A method as set forth in claim 1, wherein outputs of said voltage-fed inverters (2)(5)(12)(20) are modulated so as to output pulse signals, each pulse signal having a constant pulse width within the entire conducting interval. A method as set forth in claims 1 and 2, wherein said voltage-fed inverters (2)(5)(12)(20) are controlled so that the phase of the inverter output voltage is advanced from the phase of an induced voltage of said brushless D.C. motor (2)(5)(12)(20), the latter phase being a phase which makes a brushless D.C. motor current and the induced voltage of the brushless D.C. motor (2)(5)(12)(20) to be the same phase to one another. A method as set forth in one of claims 1 through 3, said method obtains a first voltage at a neutral point (14d) by connecting one ends of resistors (14u)(14v)(14w), the other end of each resistor (14u)(14v)(14w) is connected to each output terminal of said voltage-fed inverters (2)(5)(12)(20), obtains a second voltage at a neutral point (13d) by connecting one ends of stator windings (13u)(13v)(13w) of said brushless D.C. motor (3)(13), and detects a magnetic pole position of said rotor (3e) of said brushless D.C. motor (3)(13) based upon a difference between the first voltage and the second voltage. A method as set forth in claim 4, wherein said conducting interval of said voltage-fed inverters (2) (5) (12) (20) to be a predetermined interval which is more than 120° and less than 180° by electrical angle. A method as set forth in claim 4, wherein said conducting interval of said voltage-fed inverters (2)(5)(12)(20) to be a predetermined interval which is equal or more than 140° and equal or less than 170° by electrical angle. A brushless D.C. motor driving and controlling apparatus which drives a brushless D.C. motor (3)(13) using voltage-fed inverters (2)(5)(12)(20), said apparatus comprsing; a conducting interval determining means (18)(19) for determining a conducting interval of voltage-fed inverters (2)(5)(12)(20) to be a predetermined interval which is more than 120° and less than 180° by electrical angle, said apparatus employs a rotor (3e) which includes permanent magnets (3f) in the interior of said rotor (3e), as rotor (3e) of said brushless D.C. motor (3)(13). An apparatus as set forth in claim 7, further comprises a modulating means (18d)(18d') for modulating outputs of said voltage-fed inverters (2)(5)(12)(20) so as to output pulse signals, each pulse signal having a constant pulse width within an entire conducting interval. An apparatus as set forth in claim 7, further comprises a phase controlling means (18a)(18a')(18e) for controlling said voltage-fed inverters (2)(5)(12)(20) so that the phase of the inverter output voltage is advanced from the phase of an induced voltage of said brushless D.C. motor (3)(13), the latter phase being a phase which makes a brushless D.C. motor current and the induced voltage of said brushless D.C. motor (3)(13) to be the same phase to one another. An apparatus as set forth in one of claims 7 through 9, further comprises resistors (14u)(14v)(14w) each of which is connected its one end to each output terminal corresponding to each phase of said voltage-fed inverters (2)(5)(12)(20) and which resistors (14u)(14v)(14w) are connected their other ends to one another, a difference voltage outputing means (15) for receiving a first voltage at a neutral point (14d) which voltage is obtained by connecting the other ends of said resistors (14u)(14v)(14w), for receiving a second voltage at a neutral point (13d) which voltage is obtained at the neutral point (13d) point at which stator windings (13u)(13v)(13w) of said brushless D.C. motor (3)(13) are connected to one another, and for outputing a difference voltage between the first voltage and the second voltage, and a rotor position detecting means (16)(17) for detecting a magnetic pole position of said rotor (3a)(3e) of said brushless D.C. motor (3)(13) based upon the difference voltage. An apparatus as set forth in claim 10, wherein said conducting interval determining means (18)(19) determins a conducting interval of said voltage-fed inverters (2)(5)(12)(20) to be a predetermined interval which is more than 120° and less than 180° by electrical angle. An apparatus as set forth in claim 10, wherein said conducting interval determining means (18)(19) determinS a conducting interval of said voltage-fed inverters (2)(5)(12)(20) to be a predetermined interval which is equal or more than 140° and equal or less than 170° by electrical angle. An electrical equipment is charcterized in that said equipment employs a brushless D.C. motor (3)(13) as a driving source which motor is driven and controlled by said brushless D.C. motor driving and controlling apparatus according to one of claims 7 through 12.

Inventors: KIMURA TAIZOU, OOYAMA KAZUNOBU, YAMAI HIROYUKI, KIMURA, TAIZOU, OOYAMA, KAZUNOBU, YAMAI, HIROYUKI, KIMURA, Taizou Daikin Industries, Ltd., OOYAMA, Kazunobu Daikin Industries, Ltd., YAMAI, Hiroyuki Daikin Industries, Ltd.
Applicants: DAIKIN IND LTD, DAIKIN INDUSTRIES, LIMITED
Application year: 1995
Main classification: H02P6
IPC codes: H02P6