Multi-channel transponder with channel amplification at a common lower frequency

Abstract: Input RF signals to a transponder, such as a communications satellite, are down converted (38) to a lower intermediate frequency (IF) that is common for each different channel. The channel signals are all processed at the common IF, and then up converted (44) to different respective transmission frequencies. This allows for a down conversion/signal processing/up conversion unit (CCU) design that is the same for each channel (36) (except for frequency and gain values), and facilitates the use of inexpensive available standard components as well as low bandwidth channel filters (40). An improved phase noise cancellation phase lock loop (PLL) (48) is preferably used to generate tuning signals used in the up and down conversion processes. The PLL includes two loop mixers (66,70), with the down and up conversion mixing signals taken respectively from the inputs to the first and second loop mixers. The first and second loop mixers receive mixing signals at frequencies (R-T) and (T-IF-N·CLK) respectively, where T is the channel's transmission frequency, R is its reception frequency, N is the PLL's ÷N factor, and CLK is a clock frequency; the PLL's voltage controlled oscillator (64) provides the down conversion tuning signal and the input to the first loop mixer at a frequency (R-IF), while a (T-IF) tuning signal for the up converter is taken from a filtered output of the first loop mixer.

Description: The present invention relates to a radio frequency (RF) signal transponder receiving input signals at different respective RF reception frequencies within a first frequency band, said transponder comprising: a plurality of channel control units for processing down converted signals, and up converting the processed signals up to different respective RF transmission frequencies within a second frequency band different from the first frequency band. Such an RF signal transponder is known from US-A-4,002,980. This invention relates generally to transponder systems, and more particularly to communications transponders such as satellites which receive input signals at multiple channel frequencies, process the received signals and retransmit them at other frequencies. Satellite transponders typically receive ground-generated signals within one frequency band, such as the K u band (12.5-18 GHz), process the received signals, and then retransmit them back to earth within another frequency band such as the X-band (8-12.5 GHz). The transponder includes numerous channels that are allocated to communication traffic, with the channels separated from each other in frequency. For example, a commercial satellite will typically have 24 channels dedicated to different customers. The signals received by the satellite are processed, typically by a gain control amplifier circuit, before being retransmitted. Each channel may have different processing requirements, since each customer may have unique applications. Thus, separate gain control amplifiers are provided for each channel. The common approach currently used for satellite communications is to convert the frequency of each received signal down to its corresponding frequency for transmission back from the satellite, and to perform the gain control processing at the transmit frequency. Filtering is also performed at the transmit frequency to eliminate spurious signals. This type of system is described in Pritchard and Sciulli, Satellite Communication Systems Engineering , Prentice-Hall, Inc., 1986, pages 282-284. Performing channel filtering and gain control at the relatively high transmission frequencies requires expensive and complex microwave components to operate properly within a frequency range that may be as wide as 4-14 GHz, with a bandwidth of 500-2000 MHz. Such wide bandwidths at the transmission frequencies make it very expensive and time consuming to tune acceptable signal responses over changing temperatures and frequencies. Furthermore, discrete or at least multichip components must be used to handle the relatively high frequencies, making it very difficult to implement any of the primary functions with a single chip integrated circuit. The present approach also requires custom designed input multiplexers for each communications channel. In addition to adding to the cost and complexity of the transponder, the need for custom designed elements delays the design cycle. A satellite's frequency plan must be coordinated with all other possible sources of interference, both on the ground and in space. Because of the customized nature of present transponder architectures, satellite design and manufacturing cannot begin until permission from the Federal Communications Commission has been granted. Since the input multiplexers are custom designed, each satellite requires a mechanical design team to lay out the multiplexers and a receiver network to provide the proper interconnections and redundancy switching; this is a highly complex and expensive operation. Furthermore, switching is performed with heavy mechanical switches because of their physical location in the layout and their reliability. It would also be highly desirable to have a "frequency nimble" transponder, meaning one that is capable of easily adjusting its channel frequencies. However, the customized aspect of present transponder designs, with all filters set to fixed frequencies, does not allow for changing frequencies. For example, if a channel user finds a different market application that requires a different frequency scheme after the transponder has been designed, the customer is locked into the original frequency scheme unless it invests the substantial amount of money and time necessary for a new transponder design. Once in-orbit, no changes to the frequency scheme can ever be made. A known technique that does provide a degree of frequency nimbleness converts all of the received signals down to lower frequencies by a common down conversion factor, processes the channel signals at their separate lower frequencies, and then up converts the signals for transmission. Such an approach is described in Hughes Aircraft Company, Geosynchronous Spacecraft Case Histories, Volume II , 1981, pages iii and 2-7a.3 through 2-7a.7. However, it again requires a high degree of custom designed equipment that adds significantly to the system's cost and complexity. The signal transponder disclosed in the above-mentioned US-A-4,002,980 includes a receiver converter which transposes the complete first input frequency band down to an intermediate frequency band with identical bandwidth. Thereafter, the down converted input signal is split into a plurality of channels. In each channel, the intermediate frequency has a different value. US-A-4,972,346 discloses a high-frequency signal booster for receiving an RF signal at a certain input frequency within an input frequency band. The booster performs a down conversion to the input signal, an amplification at an intermediate frequency, and a subsequent up conversion into a frequency within the same frequency band as the input frequency band. Such a booster is particularly useful for amplifying and re-radiating high-frequency waves to a cell-drop area of a cellular mobile telephone system. In view of the above, the present invention seeks to provide a transponder system, particularly for use in satellites, that does not require custom designed components, is lighter and more compact than previous systems, eliminates the need for custom designed input multiplexers, can be implemented with integrated circuits, operates with a low noise level and is frequency nimble. Further, the present invention seeks to provide a phase noise canceling phase lock loop for use with such a signal transponder. The above objects are achieved by an RF signal transponder, as mentioned at the outset, wherein each channel control unit comprises a down converter for converting a respective input signal from within said first frequency band down to a lower common intermediate frequency (IF) which is common for all channel control units, a signal processor for processing the down converted signal at said common IF, and an up converter for converting the processed signal up to a respective RF transmission frequency. In general, the above objectives are achieved by down converting the input radio frequency signals for each channel to a lower common intermediate frequency (IF) that is the same for all channels, processing the channel signals at the common IF, and then up converting the signals to their respective transmit frequencies. Processing all of the channel signals at the same lower frequency eliminates the need for custom designed components, and allows a single common frequency to be selected for which processing components are readily and cheaply available. In the preferred embodiment, separate down conversion/signal processing/up conversion units (channel control units) are provided for each channel, with each unit including a down conversion signal mixer that is tuned to the channel's individual receive frequency to produce a signal for processing at the common IF, and an up conversion signal mixer that is tuned to the channel's transmission frequency. The tuning signals are preferably generated by said improved phase noise cancellation phase lock loop (PLL) design that includes two loop mixers, with the down conversion mixing signal taken from the input to the first loop mixer and the up conversion mixing signal taken from the input to the second loop mixer. The first and second loop mixers receive mixing signals at frequencies (R-T) and (T-IF-N·CLK), respectively, where T is the channel's transmission frequency, R is the channel's receive frequency, N is the +N factor in the PLL, and CLK is the frequency of a clock signal applied to the PLL's frequency differential detector. The PLL has a high degree of frequency nimbleness; the receive and transmit frequencies for a particular channel can be easily adjusted by changing the +N factor for that channel by a ground-based control signal, while the differential between the transmission and reception frequencies (T-R) can be adjusted by generating this mixing signal with a separate PLL that is itself adjustable from the ground. Further features and advantages of the invention will be apparent to those skilled in the art from the following detailed description, taken with the accompanying drawings. FIG. 1 is a conceptual diagram illustrating ground-based transmission to and from an orbiting satellite; FIG. 2 is a schematic diagram of an overall transponder system in accordance with the invention; and FIG. 3 is a schematic diagram of a standardized channel signal processing unit used in the transponder of FIG. 2; FIG. 4 is a perspective exploded view of inductively coupled resonator modules that are preferably used to implement the channel filters; FIG. 5 is a plan view of an assembled inductively coupled resonator pair; FIG. 6 is a circuit diagram of a cross-coupled filter that can be used for the channel filters; FIG. 7 is a plan view showing the physical structure used to implement the circuit of FIG. 6; and FIG. 8 is an enlarged fragmentary sectional view taken along the line 8-8 of FIG. 7. The basic elements of a satellite communications system to which the invention is applicable are illustrated in FIG. 1. A ground-based transmitter 2 transmits an RF signal to an orbiting communications satellite 4, which processes and retransmits the signal to a ground receiver 6. The signal from transmitter 2 may include components at numerous different channel frequencies. The satellite can also handle multiple signals originating simultaneously from different transmission locations and redirected to different corresponding receivers. The overall satellite signal processing system is shown in FIG. 2. Incoming signals are received by the satellite antenna 8 and forwarded to a band pass filter 10. The filter's pass band is typically a 500 MHz increment within the conventional 12.75-14.75 GHz satellite reception range. The filtered input signal is amplified by one of two conventional low noise amplifiers 12 and 14, which are connected in parallel in a redundancy ring formed by redundancy switches 16 and 18. Only one of the amplifiers is normally active, with the other amplifier serving as a backup in case of failure. The input signal is then divided into two equal strength signals by a hybrid signal splitter 20, with additional signal splitters 22 and 24 further splitting the divided signals into the desired channels, including redundant channels. The signal splitters can be implemented by power dividers or, in case multiple satellite antennas are employed or multiple beams are received from earth, the signal splitters could be implemented by switching networks to avoid having two different signals on the same channel at the same time. In either case, conventional signal splitters can be employed. The signal splitters divide the incoming signal into the required number of channel inputs; in the illustration of FIG. 2 sixteen different inputs 26-1 through 26-16 are used. The signal on each channel is delivered to a respective channel control unit (CCU), identified as elements CCU1-CCU16. These units receive their respective channel's portion of the input signal, process it in a unique manner that allows for a significant reduction in cost, weight and complexity while providing a high degree of frequency nimbleness, and then convert it to a frequency for transmission by the satellite. The outputs of the various channel conversion units are amplified by respective conventional high power amplifiers A1-A16 for retransmission. The amplified output signals are transmitted through a redundancy ring 28 that includes a network of redundancy switches R to replace failed channels with redundant channels, and then processed through respective isolators 28 that eliminate reflections. The isolators are connected to an output multiplexer line 30 that filters and combines the various channels for transmission back to earth by a satellite transmission antenna 34 (which may be combined with the reception antenna 8). Reject filters 32 can also be provided between the output multiplexer 20 and the transmit antenna 34. All of the elements of FIG. 2 can be implemented in a conventional manner except for the CCUs. It is a distinct advantage of the invention that, even though the various CCUs process signals are at different frequencies, a standardized CCU design is used that is substantially identical for each CCU except for the frequency values for each particular unit. This is accomplished by down converting all of the channel input signals to a lower IF which is the same for all of the different channels, performing the channel signal processing at the standardized IF, and then up converting the processed channel signals to their respective transmission frequencies. Since a frequency of 1.4 GHz is already used by many mobile cellular telephones and high quality channel controls have been developed at this frequency and are cheaply available, 1.4 GHz is a convenient IF to take advantage of the existing equipment in the cellular telephone sector. FIG. 3 shows a preferred common design for each of the conversion units CCU1-CCU16. The channel input signal on input line 36 from the signal splitter can include components at any of the channel frequencies within the satellite's reception band, which for purposes of illustration is shown as 14.0-14.5 GHz. This input signal is applied as one mixing input to a signal mixer 38 that down converts the input signal to a lower IF that is common to all of the channels. As mentioned above, this IF is preferably 1,400 MHz. The down converted signal is processed through a band pass filter 40 that is centered on the IF and has a narrow pass band. Since all of the channels down convert their respective input signals to the same lower frequency IF, the bandwidth for channel filter 40 need be only as wide as the required channel bandwidth (typically 26-72 MHz). This narrow bandwidth facilitates the use of inexpensive components with highly repeatable signal performance. The filtered IF signal is transmitted to a channel gain control element 42, which is typically formed from a network of amplifiers and variable attenuators. In general, each channel can have a different gain control requirement. The important criteria for the gain control circuit design of any particular channel include the desired channel gain (which includes the possibility of attenuation), high power efficiency, low noise, high linearity and small size and weight. The gain control element 42 replaces the channel control units or driver limiter amplifiers used in convenetional designs. Since gain control element 42 operates at an IF that is considerably lower than the high transmit frequency, an ASIC design can be used instead of the multichip or discrete designs utilized in the past. The pass band previously required was quite wide, on the order of 2 GHz, to allow a particular channel to be used as a redundancy element. The wide bandwidth and high frequency made the prior filters and gain control elements quite expensive, and also made it difficult to keep the signal performance stable over temperature. Reducing the IF to less than 2 GHz allows the gain control element 42 to be realized with silicon or gallium arsenide MMICs (monolithic microwave integrated circuits) that provide a higher circuit density and better performance and temperature stability than their prior higher frequency counterparts. The output of the gain control element 42 is applied to a second mixer 44, which up converts the signal to an output on line 46 at the channel's transmission frequency, which is illustrated as being within the transmission band of 11.7-12.2 GHz. The mixing signals used to perform both the down conversion and the up conversion are preferably obtained from a single phase lock loop (PLL) 48 of unique design. The mixing signals could also be generated with separate PLLs that are locked together, or in some applications the single-loop phase noise cancellation scheme described in U.S. Patent No. 4,918,748 to McNab et al. (corresponding to EP-A-0 315 940) could be used. The latter approach substantially eliminates the phase noise that occurs with the use of multiple PLLs, but requires the use of a filter outside the loop which must pass the up conversion mixing signal band (such as 10,300-10,800 MHz) while rejecting a nearby large local oscillator signal (typically 10,000 MHz) that is used to generate the mixing signal. Such filters are difficult and expensive to obtain. The preferred single-PLL arrangement illustrated in FIG. 3 does not require a filter that rejects large nearby signals, and thus allows for a considerable reduction in both the filter complexity and cost. The PLL 48 includes a conventional differential phase (frequency) detector 50 that receives a clock (CLK) input on line 52 and a loop return input on line 54, and produces an error signal on output line 56 that varies with any differential between the frequencies of its input signals. The error signal is processed through a loop filter 58, which is preferably preceded and followed by low pass filters 60 and 62, and controls the output of a voltage controlled oscillator (VCO) 64. The VCO is set to produce a nominal output frequency of (R-IF), where R is the channel frequency within the satellite's reception band. For example, if the channel frequency is 14,200 MHz and the IF is 1,400 MHz, the nominal VCO output would be 12,800 MHz. In FIG. 3 the VCO output is generalized as being within the range 12,600-13,100 or MHz, which covers all possible values of (R-IF) for a reception range of 14.0-14.5 GHz and an IF of 1,400 MHz. This output is applied as a tuning input to a negative input of the down conversion mixer 38 to convert the channel input signal down to the IF. An equivalent alternative would be for VCO 64 to produce an (R+IF) signal that is applied to a positive input of mixer 38, with the R signal on input line 36 applied to a negative mixer input to subtract from the (R+IF) signal from the VCO. The VCO output is also applied to a loop mixer 66, which receives a second mixing signal at a frequency of (R-T), where T is the channel's transmission frequency. The output of loop mixer 66 thus has a frequency of (T-IF), and is filtered by a pass band filter 68 whose pass band is equal to the difference between the satellite's transmission band and the IF. For example, with a transmission band of 11.7-12.2 GHz and the assumed IF of 1,400 MHz, filter 68 has a pass band of 10,300-10,800 MHz. The output of filter 68 is delivered as a tuning input to the up conversion mixer 44, to up convert the IF output of gain control element 42 to the channel transmission frequency. It is also applied to a second loop mixer 70, which receives a second mixing signal at a frequency of (T-IF-N·CLK), where N is the loop divisor factor. The output of mixer 70 thus has a frequency of N·CLK, and is transmitted through a band pass filter 72 having a pass band that passes N·CLK for all permissible values of N. The output of filter 72 is applied to a +N loop divisor 74. The range of N is equal to the satellite's reception band divided by CLK, or 500 for the illustrated example. Thus, N in this case can vary from 300 to 800, with the minimum and maximum values of N corresponding to channel reception frequencies of 14.0 and 14.5 GHz, respectively. If VCO 64 produces an (R+IF) rather than an (R-IF) output, the output of mixer 66 will be (T+IF) rather than (T-IF). In that case the IF output of the gain control element would be applied to a negative rather than a positive input of mixer 44 to produce the desired T mixer output, and a (T+IF-N·CLK) signal would be applied to the negative input to the second loop mixer 70. Other variations to the mixing signal frequencies and the polarities of the mixer inputs can be envisioned, all of which are equivalent to the particular frequency scheme shown in FIG. 3 if they produce an IF signal frequency input to the gain control element 42, and a T output frequency from mixer 44. The various frequency signals are obtained from a regulated clock source 76, which is illustrated as outputting a 50 MHz signal. This is processed by a frequency divider 78 that divides by 50 to yield the 1 MHz CLK reference input on line 52 for the differential frequency detector 50. The 50 MHz clock signal is multiplied by a factor of 46 in a first frequency multiplier 80, and by a factor of 200 in a second frequency multiplier 82, to produce the (R-T) and (T-IF-N·CLK) mixing signals that are provided respectively to the first and second loop mixers 66 and 70. The reception and transmission frequencies for each channel can be adjusted by a ground-based control signal that adjusts the +N factor for a particular channel. This commandability feature results in a frequency nimble pay load that allows the user to change the frequency plan at any time, including while the satellite is in-orbit. The PLL for a given channel can also be commanded to switch frequency schemes in case of a failure, eliminating the need for any input redundancy switches and cross connections. To adjust a channel's +N factor, a serial digital command is transmitted from the controlling ground station to the satellite's telemetry and command receiver 84, which relays the command to the satellite's command distribution unit 86. This unit directs the command to the command decoder 88 for the designated channel, which decodes the command and accesses a programmable read only memory (PROM) 90 to set a new +N value. A telemetry and command receiver 84, command distribution unit 86, and channel command decoders 88 and PROMs 90 are conventionally provided in communications satellites. Various modifications to the above described phase locked loop based synthesizer are possible. For example, the mixer 66 need not be within the loop. A fixed divider (prescaler) could be used in place of the mixer 70. A lower frequency could be used, followed by multipliers or mixers to obtain the high frequency L.O. signals. If adjustability of the separation between the receive and transmit frequencies is desired in addition to an ability to adjust these frequencies jointly, the first frequency multiplier 80 can be implemented as a separate phase lock loop having its own adjustable +N divisor element. With this dual adjustability feature, the +N element 74 of the main PLL 48 would first be adjusted to achieve a desired reception frequency, followed by a command to adjust the +N factor of the secondary PLL 80 to achieve the desired transmission frequency. Using currently available Si and GaAs ASIC and MMIC (monolithic microwave integrated circuit) technologies, many of the PLL components can be designed and manufactured on a single die. This greatly reduces the cost and improves the manufacturability of what has previously been expensive circuitry. Since the circuit designs for the PLLs of each channel are identical except for frequency, relatively large quantities of PLLs can be manufactured at one time, resulting in economies of scale that are not realizable in conventional receiver architectures which use only two to four receivers per spacecraft; the smaller quantities make it very expensive to design and manufacture PLLs for use in such units. The channel filters 40 are preferably implemented with a novel cross-ccupled ceramic filter that is small and light weight, typically less than 16cc and 85 grams. Another advantage is an excellent frequency, phase and amplitude stability over temperature, eliminating the temperature stabilizing heaters or very careful placement of the filters within the payload that conventional cavity filters require to provide a stable base temperature. Cavity filters for high frequency are also very expensive, and have reliability problems stemming from debris that is impractical to eliminate because of the nature of the manufacturing process. Ceramic filters, by contrast, do not have cavities that can trap or hold debris. Although not as preferable as the cross-coupled ceramic filter, other filters such as that described in U.S. Patent No. 4,431,977 could also be used. A preferred implementation for the channel filters 40 is shown in FIGs. 4-8. The filter is constructed from dielectric (ceramic) coaxial resonators such as the resonator described in a catalog by TransTech Corp., Coaxial Transmission Line Elements , Rev. 2, 1992, pages 1-9. Such resonators have microwave resonant frequencies. FIGs. 4 and 5 show a technique for adding shunt capacitances to a pair of dielectric resonators 102 and 104, and for inductively coupling the resonators together. The resonator is conventionally provided as a rectangular ceramic block 106 with holes 108 that extend from the top surface to the bottom surface. A conductive coating, preferably a plating 110, is deposited on the ceramic block 106 and through the hole of the block 108. Instead of the conductive plating conventionally supplied with a dielectric resonator, a preferred plating material for purposes of a microwave bandpass filter is a thick film silver-platinum ink; various combinations, including copper-silver or chrome-silver, may also be suitable. A circular ring gap 112 is formed in the conductive plating on the top surface surrounding the hole 108. The plated hole acts as a coaxial resonator, which is a short stub whose length is determined by the desired resonant frequency. At the microwave frequencies for which the invention is intended, the parameters associated with each resonator are the characteristic impedance and electrical length . To realize cross-couplings, the electrical length of each resonator is chosen to be approximately 60°. The characteristic impedance is predetermined by the dielectric constant of the material and the dimensions of the resonator itself. Although circular gaps are used for shunt capacitance, a tuning screw can be inserted into the hole of the resonator for fine tuning adjustment of the resonant frequency. The ceramic resonator 106 is exposed on one wall of the resonator by removing a section of the conductive plating 10 from the wall. In the preferred embodiment the conductive plating 10 frames the edges 114 of the coupling wall. To inductively couple the two resonators 102 and 104 together, the resonators are positioned such that their coupling walls are mated together, and a conductive shim 116 is placed between them. The shim, made of copper, can be interchanged to achieve specific inductive coupling. The shim is electrically and mechanically connected to the plating on the coupled wall 114 of each resonator. An opening 118 in the interior portion of the shim controls inductive coupling between the two resonators when they are joined together with the shim in between. The thickness of the shim and the geometry of its opening determine the degree of coupling between the two resonators. This is a very convenient mechanism that does not require any portion of the resonators to be machined away, and allows the inductive coupling between resonators to be adjusted simply by using a different shim. As explained below, capacitive couplings between different resonators are formed by connecting capacitors to the portions 120 of the resonators' conductive plating that lie within their ring-shaped gaps 112. FIG. 6 is a circuit diagram of a preferred filter configuration. It is a cross-coupled canonic design, as disclosed in Pfitzenmaier, "Synthesis and realization of narrow-band canonical microwave band pass filters exhibiting linear phase and transmission of zeros", IEEE Transactions on Microwave Theory and Techniques , Vol. MTT-30, pages 1300-1311, September 1982. The cross-coupled configuration shown in FIG. 6 has a number of advantages for applications such as a satellite transponder. It allows for the introduction of notches in the filter's response at both ends of its pass band which make its rejection steeper. It also enhances the equalization of the group delay within the pass band, helps to flatten the insertion loss and makes narrower bandwidths possible at high frequencies. A filter is shown implemented with eight resonator cells R1-R8, which consist of respective parallel capacitors CR1-CR8 and inductive transmission lines TR1-TR8 shorted to ground. Resonator pairs R1 and R8, R2 and R3, R4 and R5, and R6 and R7 are coupled inductively by T18, T23, T45 and T67, respectively. Series capacitors CO1, C12, C34, C56, C78 and C80 are inserted respectively between the input and R1, R1 and R2, R3 and R4, R5 and R6, R7 and R8, and R8 and the output. R2 and R7 are capacitively cross-coupled to each other by capacitor C27, and similarly R3 and R6 are cross-coupled together by capacitor C36. This configuration results in an eight-pole combline filter. CM1 and CM2 represent parasitic capacitances. To produce a filter with a 1 GHz center frequency and a 36 MHz pass band, all of the resonator shunt transmission lines TR1-TR8 had 13 ohm characteristic impedances and 60° electrical lengths, and the values of the other circuit elements were: CM1, CM2 - 0.19 pF C27 - 0.056 pF CR1, CR8 - 5.21 pF C36 - 0.0896 pF CR2, CR7 - 6.675 pF T23, T67 - 306 ohms CR3, CR6 - 6.80 pF T45 - 268 ohms CR4, CR5 - 6.91 pF T18 - 10135 ohms CO1, C80 - 1.51 pF C12, C78 - 0.26 pF C34, C56 - 0.274 pF . An implementation of this filter configuration with dielectric resonators is shown in FIG. 7. The physical elements which implement the various circuit functions indicated in FIG. 7 are identified by the same reference numbers as in FIG. 6. Eight dielectric resonators R1-R8 divided into two rows of four resonators each are used, although the filter could be extended to more or fewer resonators. The resonators are mounted on a common conductive collar 122 that serves as a ground plane. Filter inputs and outputs are provided by pins 124 and 126, which are respectively secured in place by a teflon insulator within brass bushings 128 and 130 above the resonator surfaces. The shorted transmission lines TR1-TR8 and the shunt capacitors CR1-CR8 represent the shunt inductance and capacitance of each resonator. To obtain the shunt inductance and capacitance values mentioned above, TransTech ceramic coaxial resonators SR8800 (High Profile) were used, with their metal plating replaced by a thick film silver-platinum ink. Each resonator as seen in FIG. 7 was 1.21×1.21 cm, while their heights were cut to 0.81 cm. The outer diameter of the ring-shaped gaps 112 was 0.965 cm for the gaps 112-1 and 112-8 of resonators R1 and R8, and 0.940 cm for gaps 112-2 through 112-7 of resonators R2-R7; the inner gap diameters were 0.81 cm for gaps 112-1 and 112-8, and 0.84 cm for gaps 112-2 through 112-7. The openings 108-1 through 108-8 of resonators R1-R8 were 0.325 cm in diameter. The inductive transmission line couplings T23, T45 and T67, and the cross-coupling transmission line T18, were implemented with respective shims 116. For inductive couplings T23 and T67 the shims had a width of 1.21 cm, a height of 0.89 cm, and an interior opening that was 0.648 cm 2 spaced 0.13 cm below the upper surface and 0.51 cm above lower surface. The width of the interior opening for the coupling T45 was extended to 0.693 cm. For the cross-coupling shim T18, a much smaller interior opening was provided to produce less coupling. In this case the shim was again 1.21 cm wide but 1.32 cm high, and the interior opening was circular with a 0.197 cm diameter centered 0.254 cm above the bottom edge. In all cases the shims were 0.0254 cm thick and made of copper. Solid shims 132 and 134 without any interior openings were used to fill in the gaps between resonator pairs R3, R6 and R2, R7. Conventional capacitors were used for capacitors C01 and C80, with one side soldered to the resonator surfaces 120-1 and 120-8 within the resonator's ring-shaped gaps, and the other side connected by ribbons 136 and 138 to the input and output pins 124 and 126, respectively. Each capacitor can be implemented in a conventional manner as a pair of parallel capacitors having opposite temperature coefficients (one positive and one negative) that provide a net temperature compensation. The other series and cross-coupling capacitors C12, C34, C56, C78, C27 and C36 are preferably constructed as shown in FIG. 8, which is an enlargement of the series capacitor C78. The same reference numbers for the various dielectric resonator elements are used as in FIG. 4, followed by a hyphen and the resonator number. A preferably solid and continuous dielectric substrate 140 is attached to the resonator conductive plating 110-7 and 110-8, spanning the resonator interface. Capacitor pads 142-7 and 142-8 are etched on the upper dielectric surface, leaving a capacitive gap 144 between the pads. The pads are the same size, with a parallel gap in between. The pads are connected to the portions of the platings 120-7 and 120-8 inside the gaps 112-7 and 112-8 by respective connector ribbons 146 and 148, which may be gold or silver. While an illustrative embodiment of the invention has been shown and described, numerous variations and alternate embodiments will occur to those skilled in the art. Such variations and alternate embodiments are contemplated, and can be made without departing from the scope of the invention as defined in the appended claims.

Claims: A radio frequency signal transponder receiving input signals at different respective RF reception frequencies within a first frequency band , said transponder comprising: a plurality of channel control units (CCUi) for processing down converted signals, and up converting the processed signals up to different respective radio frequency transmission frequencies within a second frequency band different from the first frequency band, characterized in that each channel control unit (CCUi) comprises: a down converter (38) for converting a respective input signal from within said first frequency band down to a lower common intermediate frequency, which is common for all channel control units, a signal processor (42) for processing the down converted signal at said common intermediate frequency, and an up converter (44) for converting the processed signal up to a respective radio frequency transmission frequency. The signal transponder of claim 1, characterized in that the channel control unit for each channel includes a down conversion signal mixer (38) that is tuned to a reception frequency for that channel to produce a signal at said common intermediate frequency for signal processing. The signal transponder of claim 1 or 2, characterized in that the channel control unit for each channel includes an up conversion signal mixer (44) that is tuned to a transmission frequency for that channel. The signal transponder of any of claims 1 - 3, characterized in that the channel control unit for each channel includes a phase lock loop (48) that generates tuning signals for the channel's down conversion and up conversion signal mixers (38, 44), and each channel phase lock loop (48) comprises a differential frequency detector (50), a loop filter (58), a voltage controlled oscillator (64), a first loop mixer (66), a second loop mixer (70), a ÷N frequency divider (74) connected substantially in a loop with the output of the +N frequency divider connected as a variable input (54) to the differential frequency detector (50), and a clock signal generator (76, 78) applying a clock frequency signal to a clock input (52) of said differential frequency detector (50). The signal transponder of claim 4, characterized in that a tuning signal (R-IF) for the down conversion signal mixer (38) for each channel is taken from the loop output of the voltage controlled oscillator (64) for that channel, and a tuning signal (T-IF) for the up conversion signal mixer (44) for each channel is taken from the loop input to the second loop mixer (70) for that channel. The signal transponder of claims 4 or 5, characterized in that the phase lock loop (48) for each channel includes a band pass filter (68) between its first and second loop mixers (66, 70) that is tuned to the difference (T-IF) between the transmission frequency for said channel and said common intermediate frequency. The signal transponder of any of claims 1 - 6, characterized in that said channel control units include respective amplifiers (42) for amplifying the channel signals at said common intermediate frequency, said amplifiers (42) having bandwidths substantially less than said intermediate frequency. The signal transponder of any of claims 1 - 7, characterized in that said intermediated frequency is less than 2 GHz. The signal transponder according to claim 1 comprising a phase lock loop (48) for producing a pair of output signals at respective frequencies R-IF and T-IF based upon an input clock signal frequency CLK, where a reception frequency R is preferably greater than a transmission frequency T and T is greater than an intermediate frequency IF, comprising: a signal generator (76, 80, 82) for generating a pair of mixing signals at respective frequencies R-T and T-IF-N·CLK, where N is a loop divisor factor, a differential frequency detector (50) having CLK (52) and variable frequency (54) inputs, a loop filter (58) connected to receive an output from said differential frequency detector (50), a voltage controlled oscillator (64) connected to receive an output control signal from said loop filter (58), and to produce a first loop output at said R-IF frequency, a first loop mixer (66) connected to mix said voltage controlled oscillator output R-IF with said R-T mixing signal, a second loop mixer (70) connected to mix an output T-IF from said first loop mixer (66) with said T-IF-N·CLK mixing signal, a second loop output providing said T-IF output signal at an input to said second loop mixer, and a ÷N frequency divider (74) connected to divide the frequency of an output from said second loop mixer (70) by N, and to apply the resulting frequency-divided signal to the differential frequency detector's (50) variable frequency input.

Inventors: BRUNNER ROBERT, COPELAND WILBERT B, DUNWOODY JOHN, SODERBURG ERIK, WILLIAMS KEITH, BRUNNER, ROBERT, COPELAND, WILBERT B., DUNWOODY, JOHN, SODERBURG, ERIK, WILLIAMS, KEITH
Applicants: HUGHES ELECTRONICS CORP, HUGHES ELECTRONICS CORPORATION
Application year: 1995
Main classification: H03D7
IPC codes: H04B1, H03J5, H03L7, H03D7