MICRO-POWER RAIL-TO-RAIL AMPLIFIER

Abstract: Rail-to-rail amplifier (10) has a low complexity output stage (18) that provides a high ratio of load current (IL) to no load idle current. Included are first and second output transistors (Q1, Q2) whose current circuits are connected in series at the output terminal (20) between high and low voltage supplies. Control transistor (Q3) responds to drive voltage (Vd) at its base by modulating the base-emitter voltages of the first output transistor and a gain transistor (Q4) in opposite directions to modulate their respective output and gain currents. Regenerative current source (55) supplies current to the gain transistor by returning the gain current in a regenerative feedback loop (CM1, IS1, 58) to its emitter so that the current source idles at a low gain current but is capable of supplying much higher gain currents. Voltage element (65) responds to the gain current by applying a control voltage to the base of the second output transistor so that the output transistors' output currents are unbalanced thereby generating an output current at the output terminal.

Description: Micro-Power Rail-To-Rail Amplifier BACKGROUND OF THE INVENTION Field of the Invention The present invention generally relates to bipolar micro-power rail-to-rail amplifiers, and more specifically to a micro-power output stage that is capable of delivering a high ratio of load current to no load idle current. Description of the Related Art Battery powered amplifiers may spend long periods of time in an "idle" condition with no input signal, or with intermittent input signals as in the case of voice signals. Power dissipated in these idle periods is wasted and may shorten the lifetime of the battery. The output stage of a rail-to-rail amplifier is de- signed to deliver a specified amount of signal power to a load and to consume low power during the idle periods. A standard measure for an amplifier's power consumption is its no load idle current. Present bipolar output stages contribute at least 100μA of no load idle current and re- quire a relatively large number of components to deliver the desired signal gain, suitably an open loop gain of one million and a closed loop gain of one. The amount of power dissipated during idle periods is too high for applications such as battery powered instrumentation and telecommunica- tion equipment. Furthermore, the number of components and heat dissipation requirements increase the physical dimen¬ sions of the integrated circuit, which increases its cost. Analog Devices, Inc. produces a "Precision Rail-to- Rail Input & Output Operational Amplifier" OP284 that has the ability to swing rail-to-rail at both the input and output. The OP284 has a low supply current of approximate- ly 800μ per amplifier at room temperature. The OP284's output stage uses a constant current source to supply the gain current needed to achieve sufficient signal gain, and hence idles at a relatively high current level. Analog Devices, Inc. also produces a "Micropower Sin- gle-Supply Rail-to-Rail Input/Output Op Amp" OP291. The OP291 has a low supply current of 300μ per amplifier and consumes a quiescent current of 600μ. The OP291's output stage uses output transistor that have 180x relative emit¬ ter sizes to achieve the desired gain. This requires a lot of space on the chip. Furthermore, the output stage re¬ quires a high number of components. SUMMARY OF THE INVENTION The present invention seeks to provide a bipolar mi- cro-power rail-to-rail operational amplifier having a low complexity output stage that provides a high ratio of out¬ put load current to no load idle current. This is accomplished with a dual differential input stage that converts a rail-to-rail input voltage into dual differential currents, a transimpedance stage that converts the differential currents into a single ended drive voltage at a gain node, and an output stage that amplifies changes in the drive voltage to produce a rail-to-rail output volt¬ age at an output terminal. The output stage includes first and second output transistors of opposite conductivities whose current cir¬ cuits are connected in series at the output terminal be¬ tween high and low voltage supplies. A gain transistor has a base that is held at a constant reference voltage, an emitter that is connected to the base of the npn, and a collector that conducts a gain current. A regenerative current source supplies current to the gain transistor by returning the gain current in a regenerative feedback loop to its emitter so that the current source idles at a low current but is capable of supplying much higher currents. A voltage element responds to the gain current by applying * a control voltage that varies approximately logarithmically at low gain currents and approximately linearly at high gain currents, to the base of the second output transistor so that its output current varies approximately proportion¬ al to the gain current at low levels and as an approximate¬ ly exponential function of said gain current at high lev¬ els. A control transistor having a base, a collector and an emitter that is connected to the base of said first output transistor and the emitter of said gain transistor. The control transistor responds to the drive voltage at its base by modulating the base-emitter voltages of the first output and gain transistors in opposite directions so that their output currents are unbalanced thereby generating a load current at the output terminal which adjusts the out¬ put voltage. The control transistor also conducts a con¬ trol current that stabilizes the feedback loop. For a better understanding of the invention, and to show how the same may be carried into effect, reference will now be made, by way of example, to the accompanying drawings. BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a block diagram of a micro-power rail-to- rail operational amplifier in accordance with the present invention; FIG. 2 is a simplified schematic diagram of an output stage for the operational amplifier shown in FIG. 1; and FIGs. 3a, 3b and 3c are together a schematic diagram of a preferred embodiment of the operational amplifier shown in FIG. 1. DETAILED DESCRIPTION OF THE INVENTION As shown in FIG. 1, a micro-power rail-to-rail oper¬ ational amplifier (opamp) 10 includes a transconductance input stage 12 that converts a rail-to-rail differential input voltage signal V in into dual differential currents ioi / !io " an d I h igh + ϊhigh " / a transimpedance stage 14 that converts the differential currents into a single ended drive voltage V d at a gain node 16, and an output stage 18 that amplifies changes in V d to produce a rail-to-rail out¬ put voltage V 0 at an output terminal 20. As shown, ampli¬ fier 10 is a single-sided supply amplifier having a high supply voltage V co of 5V and a low supply voltage V ee of ground potential. An external load 22 is preferably con¬ nected between output terminal 20 and V ee . The input and transimpedance stages suitably provide 60 dB of gain, and output stage 18 suitably provides another 60 dB of gain. Hence, amplifier 10, operating open loop, would produce a IV output voltage V 0 in response to a lμ differential input voltage V in . In practice, V in is applied to the input stage's non- inverting terminal 24 and output terminal 20 is connected to the input stage's inverting terminal 26 to provide nega¬ tive feedback that forces output voltage V 0 to substantially follow V in . An error voltage V e = (V in -V 0 ) of between approx¬ imately 0 and 5μV is maintained across the input terminals so that the output voltage V 0 follows V in . In this closed loop configuration, amplifier 10 is a broadband unity gain amplifier with low signal distortion that delivers a con¬ trolled amount of signal power to external load 22. Amplifier 10 is said to be in balance when it delivers zero load current. Under no load conditions, amplifier 10 can not source any load current, and hence is always bal- anced. Under load conditions, amplifier 10 is only balanc¬ ed when V 0 =V ee . During idle periods, V 0 will fall to near V ee but will not be perfectly balanced due to practical limita¬ tions in the output stage. In the preferred embodiment of the invention, amplifier 10 supplies a maximum of approxi¬ mately 5mA of load current I L to load 22 while drawing a no load idle current of only approximately 50μA so that the amplifier consumes a low amount of power during idle peri¬ ods. Input stage 12 preferably includes a pair of differ¬ ential input circuits 28 and 30, which have identical cir¬ cuit topologies except that their transistor conductivities are reversed. Input circuit 28 operates for input voltages V in down to V ee , but shuts down when V in gets to within two diode voltages of a high supply voltage V cc . Similarly, in¬ put circuit 30 operates for input voltages V in up to V cc but shuts down when V in gets to within two diode voltages of V ee . Together input circuits 28 and 30 allow V in to swing between approximately the low and high supply voltages V ee and V cc , respectively, without clipping. Both input circuits are turned on over a majority of the input signal range and one of the two are turned on when V in approaches one of the sup¬ ply voltages. Alternately, the input circuits could be controlled so that only one was turned on at a time. Input circuits 28 and 30 produce differential currents ϊiov I l o " and I high + Ihigh " - Under no load conditions, ampli¬ fier 10 does not supply load current at output terminal 20 and is said to be in balance so that these currents are equal. Under loaded conditions, the differential currents are imbalanced to supply load current I L at terminal 20 and maintain V 0 at substantially V in . When driving load 22, input circuits 28 and 30 respond to changes in V in , and hence V e by adjusting differential currents I low + and I high + so that they move in opposite directions and adjusting I low " and h i gh " so that they move in opposite directions to modulate the load current supplied at terminal 20. The negative feedback maintains the differential currents at approxi¬ mately their no load values. Transimpedance stage 14 responds to changes in the differential currents produced by input stage 12 by reduc¬ ing drive voltage V d over a range of approximately 0 to 5mV relative to a known reference voltage to produce a drive current at gain node 16. Input circuit 28 and a high side of transimpedance stage 14 are tied through resistors Rl and R2 to V cc to supply differential currents I low \ I low " to input circuit 28 and currents I l r I 2 to the transimpedance stage. Similarly, input circuit 30 and a low side of transimpedance stage 14 are tied through resistors R3 and R4 to V ee to draw differential currents I high + , I h i gh " from in- put circuit 30 and currents I 3 , I 4 from the transimpedance stage. Under no load conditions, currents I x through I 4 are substantially the same and drive voltage V d remains at the reference voltage so that the drive current at gain node 16 remains constant. Under loaded conditions, the differen- tial currents are imbalanced which reduces drive voltage V d and the amount of drive current supplied at gain node 16. Transimpedance stage 14 is designed so that the vol¬ tages across resistors Rl and R2 are substantially con¬ stant. Therefore, changes in I low + and I low " induce equal and opposite changes in I x and I 2 . Transimpedance stage 14 for¬ ces I 3 to be substantially equal to I 1 so that changes in I low + and I high + adjust the voltage across resistor R3 to re¬ flect the change in V in . Transimpedance stage 14 is further designed so that the voltage across resistor R4 tries to follow the voltage across R3. This causes currents I 2 and I 4 to move in oppo¬ site directions in response to a change in V in thereby mod¬ ulating drive voltage V d relative to the reference voltage and adjusting the drive current at gain node 16. Drive voltage V d is preferably applied to a buffer circuit 32, which shifts V d towards the low supply V ee by an amount substantially equal to the nominal voltage across resistor R3. This is done to set the value of V d at balance to the particular reference voltage level and to reduce the amplifier's offset voltage. Circuit 32 also buffers the transimpedance stage so that it does not have to directly supply the drive current to output stage 18. Output stage 18 responds to the level shifted and buf¬ fered voltage signal, also referred to as V d by supplying load current I L at output terminal 20 to produce output voltage signal V 0 across external load 22. Output stage 18 uses a regenerative current source to provide an internal gain current that is used to control signal gain. The re¬ generative current source supplies the gain current in a regenerative feedback loop so that the gain current can vary over a wide range, approximately OμA to approximately 110μA, in response to changes in drive voltage V d . This allows output stage 18 to supply the large amount of gain current needed to provide a high gain output stage while idling under no load conditions at approximately 3μA. Un¬ der loaded conditions, the regenerative current source will idle at a somewhat higher level because in practice output stage 18 can not be driven all the way to ground, and hence a small current will flow through the load. By supplying gain current as it is demanded by changes in V in , output stage 18 lowers the average power consumed by the amplifier without sacrificing performance. In the pre¬ ferred embodiment amplifier 10 consumes a total no load idle current of approximately 50μA with only approximately 8μA being consumed by the output stage. The maximum load current is 5mA so that the ratio of load current to no load idle current is approximately one thousand. Furthermore, the regenerative current source is self-limiting, which prevents the gain current from increasing unchecked in re- sponse to an extreme variation in V d and destroying the amplifier. The regenerative current source requires rela¬ tively few components and a moderate amount of chip space. FIG. 2 is a schematic diagram of output stage 18 shown in FIG. 1. Output stage 18 comprises a lOx pnp output transistor Ql whose emitter 34 is connected to V cc and a 4x npn output transistor Q2 whose emitter 36 is connected to V ee . Their collectors 38 and 40 are connected together at output terminal 20. The lOx and 4x designations represent the transistors emitter sizes relative to standard lx size, and are chosen to provide proper bias conditions for the amplifier. In general, current flows from the supply V cc through transistors Ql and Q2 and into negative supply V ee . Under no load conditions, the collector currents I Q1 and I Q2 are equal, suitably 4μA, so that amplifier 10 neither sinks nor supplies a difference current I L at output terminal 20. Under load conditions, output stage 18 responds to V in by imbalancing the collector currents so that together tran¬ sistors Ql and Q2 source load current I L to load 22 so that V 0 follows V in . Alternately, if load 22 was connected to V cc instead of V ee , transistors Ql and Q2 would sink current from the load. Drive voltage V d is applied to the base 42 of a lx npn control transistor Q3 whose collector 44 is connected to V cc and conducts a control current I c and whose emitter 46 is connected to the base 48 of output transistor Q2. Transis¬ tors Q2 and Q3 set the reference voltage for V d at two base- emitter voltages above V ee . Control transistor Q3 responds to variations in V d by adjusting the voltage at base 48 of transistor Q2, which changes its collector current I Q2 . This imbalances the output transistors' collector currents and adjusts load current I L . Because the change in voltage at base 48 of transistor Q2 is the same as the change in drive voltage V d , the gain provided by transistor Q2 alone is relatively small. A lx npn gain transistor Q4 has an emitter 50 that is connected to the emitter 46 of control transistor Q3 for the purpose of sensing changes in V d . A voltage source V ref clamps the voltage at base 52 of transistor Q4 at two base- emitter voltages above V ee so that at balance V d and V ref are the same and under load conditions V d is less than V ref . Therefore, gain transistor Q4 and output transistor Q2 re¬ spond to a change in V d by adjusting their base-emitter voltages by equal but opposite amounts. Transistor Q4 re¬ sponds to changes in its base-emitter voltage by adjusting a gain current I g that flows through its collector 54. A regenerative current source 55 includes a current mirror CMl that mirrors gain I g from collector 54 of gain transistor Q4 to a current limiter 58, which in turn sup¬ plies current I sup to emitter 50 of gain transistor Q4. Current mirror CMl includes a diode Dl that is connected across the base-emitter junction of a lx pnp transistor Q5. The base/collector 56 of diode Dl is connected to collector 54, and the collector 57 of transistor Q5 is connected to current limiter 58. A constant current source IS1 injects current I sl , suitably lμA, into the signal path at collector 57 to increase the amount of current I lim = (I g +I sl ) that is supplied to current limiter 58. During normal operation, current limiter 58 multiplies the current I lim by a constant factor greater than one, suitably 1.5, and supplies it as current I sup to emitters 50 and 40 of gain and control tran¬ sistors Q4 and Q3. If drive voltage V d is reduced too much, control tran¬ sistor Q3 turns off so that all of the regenerated current is supplied to gain transistor Q4. Because the signal path has a gain greater than one, gain current I g if left un¬ checked would increase without bound and destroy the ampli¬ fier. Therefore, current limiter 58 responds to large gain currents I g by reducing its gain and eventually shutting down to prevent the amplifier from destroying itself. Current limiter 58 is preferably a current peaking circuit that includes a lx npn transistor Q6 having an emitter 59 that is tied to V ee , a base 60 that is connected to collector 57 of transistor Q5, and a collector 61. A resistor R5, suitably 100 ohms, is connected between base 60 and collector 61. A 1.5x npn transistor Q7 has an emit¬ ter 62 that is connected to V ee , a base 63 that is tied to collector 61 of transistor Q6, and a collector 64 that is returned to emitter 50 of gain transistor Q4. When gain current I g is small the voltage across resistor R5 is very small so that transistors Q6 and Q7 act like a current mir¬ ror with a gain of 1.5 to supply I g to gain transistor Q4. As I g increases, the base-emitter voltage of transistor Q6 increases logarithmically which tends to cause transistor Q7 to increase I sup . However, the voltage across resistor R5 increases linearly and drives the voltage at collector 61 of transistor Q6 towards V ee which tends to reduce I sup . Thus, current limiter 58 begins to limit at the point where the voltage across resistor R5 increases faster than the base-emitter voltage of transistor Q6. This reduces the base voltage at transistor Q7, which limits current I g . The regenerative feedback loop is described by a pair of equations that must be satisfied to stabilize output stage 18. The currents summed at emitter 46 of control transistor Q3, ignoring the base current of transistor Q2 and assuming that the emitter currents of transistors Q3 and Q4 equal their respective collector currents I c and I g , satisfy: The current I sup supplied by current limiter 58 is described by the following feedback equation: I SUP = G(I + I S1 ) (2) where G is the gain of current limiter 58, suitably 1.5 during normal operation and progressively smaller as lim¬ iting occurs. At balance, current limiter 58 preferably supplies 3μA of current (I sup =3μA), which is split between transistors Q3 and Q4 so that I c =2μA and I g =lμA. These values satisfy both equations 1 and 2 so that amplifier 10 is stable. Under load conditions, gain transistor Q4 responds to chan¬ ges in its base-emitter voltage by adjusting I g . Regenera- tive current source 55 senses the change in I g and adjusts I sup to supply the required gain current. Control transistor Q3 absorbs any excess current supplied by the regenerative process by adjusting I c so that equations 1 and 2 are again satisfied. A current controlled voltage element 65 is connected between current mirror CMl and V cc so that the current (2I g ) drawn from the positive supply to supply current mirror CMl flows through voltage element 65. Thus, twice the gain current I g modulates a control voltage V c at the base 66 of output transistor Ql. Control voltage V c moves in the same direction as the voltage applied to base 48 of output tran¬ sistor Q2. Therefore, transistor Ql adjusts its collector current I Q1 in opposition to the change in collector current I Q2 , which further imbalances output stage 18 and increases drive current I L . Voltage element 65 includes a diode D2, which is im¬ plemented as a diode connected 5x pnp transistor, whose base/collector 67 is connected to base 66 of output tran¬ sistor Ql and the input of current mirror CMl. A resistor R6, suitably 2.9K, is connected between V cc and D2's emitter 68. Thus, the base-emitter voltage across output transis¬ tor Ql is the sum of the voltages across resistor R6 and diode D2, which increase linearly and logarithmically, re¬ spectively, as gain current I g increases. At balance, the voltages across resistor R6 and diode D2 are approximately 5.8mV and 0.7V, respectively. There¬ fore, the diode voltage dominates and the resistor voltage can be ignored. As a result, voltage element 65 mirrors I g with a fixed amount of gain to output transistor Ql so that its collector current I Q1 is proportional to gain current I g . However, as gain current I g increases the voltage drop across resistor R6 becomes progressively more significant, and at some point the change in control voltage V c becomes approximately linear. Output transistor Ql responds to a linear change in base-emitter voltage by adjusting its col¬ lector current I Q1 as an approximately exponential function of gain current I g . The effect of this is that output transistor Ql sourc¬ es a lot more current to load 22 for a given change in drive voltage V d than output transistor Q2 sinks for an equivalent opposite change in V d . Thus, output stage 18 re¬ sponds asymmetrically to changes in drive voltage V d and only provides high gain when sourcing current to the load. Thus, the load is preferably connected to low supply V ee so that output stage 18 never sinks current. Alternately, the high gain output transistor could be used to sink current, in which case the load would be connected to V cc . In prac¬ tice, the nonlinear gain does not matter unless the ampli¬ fier is being used for a special application that requires gain linearity. Furthermore, the benefits of reduced no load idle current outweigh the minimal effects of gain non- linearity. FIGs. 3a, 3b and 3c are together a schematic diagram of a preferred embodiment of amplifier 10 shown in FIG. 1. The preferred input stage 12 is similar to those used in operational amplifiers OP-284 and OP-291 produced by Analog ^Devices, Inc., the assignee of the present invention. In¬ put circuit 28 comprises a pair of 2x pnp transistors Q8 and Q9 whose bases 69 and 70 provide the non-inverting and inverting inputs 24 and 26, respectively. Their collectors 72 and 74 are connected to V ee . A pair of 2x npn transis¬ tors QIO and Qll have bases 76 and 78 that are tied togeth¬ er at node 80, and emitters 82 and 84 that are connected to the respective emitters 86 and 87 of transistors Q8 and Q9. The collectors 88 and 89 of transistors QIO and Qll are connected to resistors Rl and R2 and conduct input currents I low + and I low " , respectively. A pair of diodes D3 and D4, which are implemented as diode connected lx npn transistors, are connected between node 80 and respective emitters 82 and 84 of transistors QIO and Qll. Thus, currents I low + and I low " in transistors QIO and Qll are twice the currents in D3 and D4, respectively. A current source IS2 supplies current I s2 suitably 1.5μA, to node 80 to supply transistors QIO and Qll, and diodes D3 and D4. At balance, supply current I s2 is split equally be¬ tween the two sides of differential input circuit 28 so that I low + and I low " conduct 1.5μA. Input circuit 28 responds to a relative increase in V in at the base 69 of transistor Q8 (non-inverting terminal 24) by reducing and increasing the voltages across diodes D3 and D4, respectively, so that a smaller portion of I s2 flows through diode D3 and a larger portion flows through diode D4. The changes in diode cur¬ rent are reflected into I low + and Iι ow " . The topology of input circuit 28 allows V in to swing from the low supply voltage V ee to within two base-emitter voltages of positive supply voltage V cc . The base-collector voltages of transistors Q8 and Q9 can operate effectively with zero volts. Therefore, V in can include V ee without turning input circuit 28 off. Conversely, if V in rises to within two base-emitter voltages of V cc , the transistors begin turning off. The input stage's dual input circuits 28 and 30 are identical except for the reversed conductivities of their transistors and diodes, and thus a detailed description of input circuit 30 is omitted. In general, input circuit 30 includes a pair of npn transistors Q12 and Q13 whose bases are tied to the non-inverting and inverting terminals, re¬ spectively. A current source IS3 draws a reference current from a pair of diodes D5 and D6. The diodes are connected across the base-emitter junctions of pnp transistors Q14 and Q15, and respond to changes in V in to modulate their collector currents I high + and I high " which flow through resis¬ tors R3 and R4, respectively. The preferred transimpedance stage 14 is the folded cascode circuit used in opamp OP-284 produced by Analog Devices, Inc. The folded cascode circuit comprises a pair of 2x pnp transistors Q16 and Q17 having bases 90 and 92 that are connected together at node 94, and emitters 96 and 98 that are connected to resistors Rl and R2 and conduct currents I x and I 2 , respectively. A current source IS4 draws a bias current I s4 , suitably 0.5μA, from node 94. A pair of 2x npn transistors Q18 and Q19 have bases 100 and 102 that are connected together at node 104, and emitters 106 and 108 that are connected to resistors R3 and R4 and conduct currents I 3 and I 4 , respectively. The collectors 110 and 112 of transistors Q16 and Q18 are connected to¬ gether and conduct the same current, 1 3 =1 ! . The collectors 114 and 116 of transistors Q17 and Q19 are connected at gain node 16 and together source current at gain node 16 in response to changes in V in . The folded cascode circuit establishes a constant voltage drop across resistors Rl and R2, suitably 50K, by fixing the voltage at node 94. A resistor R7, suitably 45K, is connected between V cc and the emitter 118 of a diode D7, which is implemented as a diode connected 1.5x pnp transistor, whose base/collector 120 is connected to node 94. Diode D7 and resistor R7 are biased so that the volt¬ ages across resistors Rl and R2 are approximately the same as the voltage across R7. As a result, 4.5μA of continuous current flows through resistors Rl and R2. The folded cascode circuit uses a lx npn transistor Q20 to control transistor Q18 and force I 3 to be sub¬ stantially the same as I l The base 122 of transistor Q20 is connected to collector 110, its collector 124 is con¬ nected to node 94, and its emitter 126 is connected to node 104. If I 3 falls below I 1# current is driven into base 122 of Q20. This pulls up base 100 of Q18 so that it increases I 3 . Conversely, if I 3 rises above I 17 the base drive to Q20 is reduced. This pulls base 100 of Q18 down, which in turn reduces I 3 . The result is that the folded cascode circuit drives current I 3 through resistor R3, which together with current I high + from input circuit 30 set the voltage across R3. Cur- rents I 3 and I high + respond to changes in V in by moving in the same direction so that the voltage across R3 reflects that change. The voltage at base 100 of Q18, and hence the voltage at base 102 of Q19 move up and down with the volt¬ age across R3 to keep I 3 the same as I x . Transistor Q19 re- sponds to variations of its base voltage by adjusting I 4 so that the voltage across R4 tries to follow the voltage across R3. At balance, I 1 = I 2 = 3 μA and I high + = I h i gh " = 1.5μA. Since the folded cascode circuit forces I 3 to equal I x , I 3 is also 3μA. Thus, transistor Q19 supplies I 4 at 3 μA so that the voltages across resistors R3 and R4 are equal. Because currents I 2 and I 4 are equal, drive voltage V d at gain node 16 remains at its reference value and the drive current provided to the next stage is unchanged. Under load conditions, amplifier 10 reduces V d relative to its reference voltage so that output stage supplies load current to hold V 0 at V in . Amplifier 10 responds to an in¬ crease in V in by increasing I high + and I 1# and hence I 3 , and reducing I 2 and I h i 3h " ■ This increases the voltage across resistor R3, which in turn increases the voltage at base 102 of transistor Q19. As the voltage across resistor R3 is increased, the voltage across resistor R4 that is at¬ tributable to I hιgh " is reduced. Thus, transistor Q19 in¬ creases I 4 to pull up the voltage across resistor R4. This pulls the voltage at its collector 116 and at gain node 16 down thereby reducing drive voltage V d and reducing the drive current. Amplifier 10 responds to a reduction in V ιn by increasing drive voltage V d and increasing the drive cur¬ rent at gain node 16. Buffer circuit 32 is connected between gain node 16 and base 42 of control transistor Q3 in output stage 18. The primary purpose of buffer circuit 32 is to establish a voltage drop equal to the voltage across resistor R3 at balance between collector 116 of transistor Q19 and base 42 of control transistor Q3. This keeps the voltages at the collectors 112 and 116 of transistors Q18 and Q19 approxi¬ mately equal, which lowers the amplifier's offset voltage. A secondary purpose is to buffer gain node 16 from base 42 of control transistor Q3 so that the change in base current does not have to be provided entirely by transimpedance stage 14. Buffer circuit 32 includes a lx npn transistor Q21 having a base 128 that is connected to gain node 16, a col¬ lector 130, and an emitter 132 that is connected to one side of a resistor R8. A diode D8, implemented as a lx diode connected npn transistor, has a base/collector 134 that is connected to base 42 of control transistor Q3 and an emitter 136 that is connected to the other side of re¬ sistor R8. Transistor Q21 and diode D8 are biased so that the voltage drop across resistor R8 is approximately equal to the voltage across resistor R3 at balance. A current mirror CM2 mirrors the collector current in transistor Q21 to the base/collector 134 of diode D8 to provide the base current that drives control transistor Q3. Therefore, a change in base current at transistor Q3 of Δl only requires a change in drive current at gain node 16 of —^- where β is the current gain of transistor Q21. Current mirror CM2 includes a diode D9, implemented as a diode con¬ nected lx pnp transistor, and a lx pnp transistor Q22. The emitters 138 and 140 of diode D9 and transistor Q22, re¬ spectively, are connected to resistor R7 to draw supply current from V cc . (This also provides the bias current that sets the voltage across resistor R7) . The base/collector 142 of diode D9 is connected to the collector 130 of tran- sistor Q21 and to the base 144 of transistor Q22. Transis¬ tor Q22's collector 146 is connected to base/collector 134 of diode D8. As described above in FIG. 2, output stage 18 responds to drive signal V d by supplying load current I L at output terminal 20 to an external load thereby modulating its out¬ put voltage V 0 between ground and 5V. Regenerative current source 55 supplies gain current I g as needed to respond to changes in drive voltage V d and provide the desired gain. When amplifier 10 is idling, either because input voltage signal V in is not being applied or because the signal pauses momentarily, regenerative current source 55 supplies only a minimal amount of gain current. This lowers the power consumption of amplifier 10 without sacrificing perform¬ ance. This saves money, lengthens the life of amplifier 10, and reduces the physical dimensions of the amplifier IC. Amplifier 10 uses a single bias circuit 148 to estab¬ lish the reference voltage level at the base 52 of transis¬ tor Q4 in output stage 18, supply current to buffer circuit 32 and resistor R7, and supply current for diode D7. This is done to conserve current and lower the amplifier's power consumption during idle periods. A current source IS5 supplies bias current I s5 , suit¬ ably 2μA, that flows from high supply V oc , through 2x diodes D10 and Dll and into V ee . The base/collector 150 of diode D10 is connected to base 52 of transistor Q4. This sets the base voltage at two base-emitter voltages above the low supply V ee . The base/collector 152 of diode Dll is connect- ed to the bases 154 and 156 of transistors Q23 and Q24, respectively. Their emitters 158 and 160 are connected to V Transistor Q23 has an emitter area of 3x, and thus supplies 3μA of current. This current is split equally between transistor Q21 and diode D8. Transistor Q24 has an emitter area of 1.5x, and thus supplies 1.5μA of current that is drawn from high supply V cc through resistor R7, di¬ ode D7, transistor Q20, and into its collector 162. Amplifier 10 conserves current and thus lowers power consumption by recycling the bias currents for use in mul¬ tiple parts of the amplifier. However, sustained extreme inputs could cause amplifier 10 to self-destruct. To pre¬ vent this, a pair of clamps QCl and QC2 are added to the folded cascode circuit, and a current limiter QL1 is added to output stage 18. Clamp QCl is a lx npn transistor having a base 164 that is connected to transistor Q18's collector 112, a col¬ lector 166 that is connected to V cc , and an emitter 168 that is connected to transistor Q19's collector 116. In normal operation, the voltages at collectors 112 and 116 are ap¬ proximately the same so that QCl is turned off. However, if transistor Q19's collector voltage drops too much, QCl turns on, which pulls Q19 up and prevents it from saturat¬ ing. Clamp QC2 is a lx npn transistor having a base 170 that is connected to emitter 136 of diode D8, a collector 172 that is tied to V cc , and an emitter 174 that is con¬ nected to collector 162 of transistor Q24. In normal oper¬ ation QC2 is turned off. However, if transistor Q17's voltage pulls up too much, it may saturate. This pulls up base 170 of clamp QC2, which turns it on. Thus, the cur¬ rent supplied by transistor Q24 is provided to clamp QC2 instead of diode D7. This turns the folded cascode circuit off until input voltage V in is reduced. Current limiter QL1 includes a lx pnp transistor Q25 having a base 176 that is connected to collector 44 of con¬ trol transistor Q3, an emitter 178 that is tied to V cc , and a collector 180 that is connected to collector 130 of buff¬ er transistor Q21. A resistor R9 is connected across the base-emitter junction of transistor Q25. In normal opera¬ tion, transistor Q25 is off. However, if drive voltage V d , and hence the voltage at base 42 of control transistor Q3 is increased too much, transistor Q25 will turn on and sup¬ ply the collector current for transistor Q21. This reduces the current in current mirror CM2 which in turn reduces the base drive to transistor Q3 and increases its base voltage. Amplifier 10 uses four capacitors C1-C4 to control its frequency response. Capacitor Cl, suitably 15pF, is con¬ nected between output terminal 20 and gain node 16. Cl is the primary compensation capacitor and serves three func¬ tions. First, Cl provides local negative feedback from the output of the amplifier back to base 42 of control transis¬ tor Q3. Cl also provides pole splitting that drives the low and high frequency poles of the amplifier outside its frequency range. Finally, Cl controls the amplifier's slew rate. Capacitor C2 is connected from transistor Q18's col¬ lector 112 to V ee . C2 reduces the high frequency transcon- ductance of the folded cascode circuit and limits the am¬ plifier's bandwidth. Capacitor C3 is connected between gain node 16 and base/collector 134 of diode D8. C3 stabi¬ lizes the amplifier by compensating for the parasitic ca¬ pacitance at the collector of Q23. Capacitor C4 is con¬ nected between gain node 16 and emitter 46 of control tran¬ sistor Q3. C4 prevents the amplifier from oscillating. While several illustrative embodiments of the inven- tion have been shown and described, numerous variations and alternate embodiment will occur to those skilled in the art. For example, different input and transimpedance stag¬ es could be used with the output stage of the present in¬ vention. Such variations and alternate embodiments are contemplated, and can be made without departing from the spirit and scope of the invention as defined in the append¬ ed claims.

Claims: I CLAIM : 1. An amplifier output stage, comprising: first and second supply voltage terminals (V cc , V ee ) ; an output terminal (20) for connection to an ex¬ ternal load (22) ; first and second output transistors (Ql, Q2) of opposite conductivities having bases (66,48) , collectors (38,40) that are connected at said output terminal and emitters (34,36)) that are respectively connected to said first and second supply voltage terminals so that said transistors conduct respective output currents (I Q1 ,I Q2 ) that produce a load current (I L ) at said output terminal; a gain transistor (Q4) having a base (52) that is held at a reference voltage, an emitter (50) and a collec¬ tor (54) that conducts a gain current (I g ) that varies over a predetermined range; a regenerative current source (55) that supplies current to said gain transistor by returning said gain cur- rent in a regenerative feedback path (CMl, IS1, 58) to its emitter so that the current source supplies a no load idle current near a lower end of said predetermined range and is capable of supplying gain current over said range; a voltage element (65) that responds to said gain current by applying a control voltage to the base of the second output transistor so that its output current varies as an approximately exponential function of said gain cur¬ rent; a control transistor (Q3) having a base (42) , a collector (44) and an emitter (46) that is connected to the base of said first output transistor and the emitter of said gain transistor and conducts a control current (I c ) that stabilizes the feedback path, said control transistor responding to a drive voltage signal (V d ) applied to its base by modulating the base-emitter voltages of said first output and gain transistors in opposite directions so that said output currents are unbalanced thereby generating said load current at said output terminal. 2. The output stage of claim 1, wherein said voltage element comprises: a resistor (R6) ; and a diode (D2) that is connected in series with said resistor between said second supply voltage terminal and the base of said second output transistor so that a bias current that is proportional to said gain current flows therethrough and sets the control voltage as the sum of the voltages across the resistor and the diode, said voltage element responding to gain currents in a lower portion of said range by modulating the control voltage approximately logarithmically so that said second transistor's output current is approximately proportional to said gain current and responding to gain currents in an upper portion of said range by modulating the control volt¬ age approximately linearly so that said second transistor's output current is approximately said exponential function of said gain current. 3. The output stage of claims 1 or 2, wherein said regenerative current source comprises: a current mirror (Cml) having a first side that is connected to the collector of said gain transistor and mirrors said gain current to a second side, and an input that draws approximately two times said gain current from said voltage element; a current source (IS1) that adds a bias current to the gain current at the second side of the current mir- ror; and a current limiting circuit (58) that is connected between the second side of the current mirror and the emit- ter of said gain transistor, said current limiting circuit responding to an increase in gain current by progressively reducing the gain of said regenerative feedback path so that the gain current supplied by the regenerative current source is limited to a maximum value. 4. The output stage of claim 3, wherein said current limiting circuit comprises: a limiting transistor (Q6) having a base that is connected to the second side of the current mirror, a col- lector and an emitter that is connected to the first supply voltage terminal; a supply transistor (Q7) having a base that is connected to the limiting transistor's collector, a collec¬ tor that is connected to the emitter of said gain transis- tor, and an emitter that is connected to the first supply voltage terminal; and a limiting resistor (R5) that is connected be¬ tween the base and the collector of said limiting transis¬ tor so that as said gain current increases the voltage at the base of said supply transistor is driven towards the first supply voltage which causes the supply transistor to limit the current supplied to the gain and control transis¬ tors. 5. The output stage of claim 4, wherein said limit¬ ing and supply transistors have emitter areas, the emitter area of said supply transistor being larger than that of said limiting transistor. 6. The output stage of claims 1,2 or 3, further com¬ prising a capacitor (Cl) that is connected between said output terminal and the base of the control transistor to provide negative feedback to stabilize the output stage. 7. A method for controlling a bipolar output stage (18) that comprises first and second output transistors (Q1/Q2) of opposite conductivities having bases, emitters that are respectively connected to first and second supply voltage terminals (V cc ,V ee ) , and collectors that are connect¬ ed at an output terminal (20) and conduct respective output currents, the method comprising: connecting a load (22) between said output termi¬ nal and said first supply voltage terminal; applying a drive voltage signal to the base of said first output transistor to adjust its output current thereby imbalancing said output currents and producing a load current (I L ) at said output terminal; generating a gain current (I g ) in response to said drive voltage signal; supplying said gain current by feeding it back in a regenerative feedback path so that said gain current can vary over a predetermined range while idling at a lower end of said range; and modulating a control voltage (V c ) at the base of said second output transistor as an approximately linear function of said gain current so that its output current is approximately an exponential function of said gain current thereby further imbalancing said output currents to in¬ crease the gain of said load current. 8. The method of claim 7, wherein supplying said gain current comprises: mirroring (CMl) said gain current into said re¬ generative feedback path; injecting current (IS1) into said regenerative feedback path to increase said gain current; returning (58) said increased gain current to supply said gain current; and adjusting a control current to absorb excess gain current and stabilize said regenerative feedback path. 9. The method of claims 7 or 8, wherein returning said increased gain current comprises: multiplying said increased gain current by a de¬ sired gain factor, said gain factor having a value greater than one for gain currents at a lower end of said predeter¬ mined range; and progressively reducing said gain factor as said gain current increases thereby limiting the amount of gain current supplied to less than a maximum value. 10. The method of claims 7,8 or 9, wherein modulating said control voltage comprises: adjusting a first voltage as a logarithmic func¬ tion of said gain current; adjusting a second voltage as a linear function of said gain current; summing said first and second voltages to provide said control voltage at the base of said second output transistor so that said control voltage changes approxi- mately logarithmically for gain current at a lower end of said range and changes progressively more linearly as said gain current increases.

Inventors: BUTLER JAMES R, BUTLER, JAMES, R.
Applicants: ANALOG DEVICES INC, ANALOG DEVICES, INC. (ADI)
Application year: 1995
Main classification: H03F3
IPC codes: H03F3