PHASE SHIFTING DEVICE

Abstract: A circuit having nearly equally halves inputted electric power in such a way that the difference between the phases of the two outputs is 90°. The outputs are fed to a first exciter which generates a circularly polarized wave. The wave is inputted into a second exciter, and the two outputs of which is combined in a synthesizing circuit. The opened ends of a shield case housing the first exciter and another shield case housing the second exciter are fitted together, and the first and second exciters, together with the cases, can be rotated relatively around the axis of the exciters. The phase difference between the input of the halving circuit and the output of the synthesizing circuit changes with the relative rotational angle between the first and second exciters.

Description: TECHNICAL FIELD The present invention relates to a phase shifting device used in, for example, a transmitting device capable of high output through parallel operation of a plurality of transmitters and a power supply apparatus for directional control of antennas. BACKGROUND ART When attempting to vary the direction of maximum radiation or side lobe characteristic of an array antenna by appropriately controlling the driving phase of each of a plurality of radiating elements forming the array antenna, or when attempting to realize a high-output transmitting device by parallel-operating a plurality of transmitters and controlling the output phase of each transmitter such that each is the same phase, line stretchers constructed so as to mechanically vary the length of the transmission line of a signal, or waveguide-type phase shifting devices formed by inserting a dielectric plate within a waveguide have conventionally been used. Fig. 1 presents several views of a waveguide-type phase shifting device of the prior art, Fig. 1A showing a plan view, Fig. 1B showing a side view, Fig. 1C showing a sectional view taken at the B-B line of Fig. 1B, and Fig. 1D showing a sectional view taken at the A-A line of Fig. 1A. In rectangular waveguide 11, flanges 12 1 and 12 2 are provided for inserting and connecting such a waveguide-type phase shifting device within a rectangular waveguide circuit. A dielectric plate 13 is provided within rectangular waveguide 11 such that the plate surface is parallel to the electric field within the waveguide 11. As shown in Fig. 1D, the contour of this dielectric plate 13 is formed as a parallelogram, this parallelogram having inclines on the edges of the radio-wave incident side and edges of the opposite side which are formed so as to improve the radio-wave reflection characteristic at these edge portions. Instead of providing inclines to the edges of the radio-wave incident side and edges of the opposite side of this dielectric 13, the radio-wave incident portion and opposite side portion of the dielectric 13 may be formed such that the plate thickness gradually varies. Support fittings 14 1 and 14 2 for dielectric plate 13 pass through the opposing short sides of rectangular waveguide 11 and dielectric plate 13. Support fittings 14 1 and 14 2 may freely slide at the portions where they pass through the opposing short sides of rectangular waveguide 11 but are secured to the portions where they pass through dielectric plate 13. Coupling plate 15 links together support fittings 14 1 and 14 2 . This coupling plate 15 serves as a handle for moving support fittings 14 1 and 14 2 either forward or backward in the axial direction of each of support fittings 14 1 and 14 2 while maintaining dielectric plate 13 in the attitude shown in Fig. 1D, thus allowing the plate surfaces of dielectric plate 13 to be moved from a position coinciding with the central axis in the longitudinal direction of rectangular waveguide 11 to a position close to either of the opposing short sides as shown in Fig. 1C, thus enabling variation in the proportion of shift change of radio waves propagated through rectangular waveguide 11. In other words, such a phase shifting device uses the change in propagation speed of radio waves within rectangular waveguide 11 according to the dielectric constant, thickness, and length in the direction of wave propagation of dielectric plate 13. When interposed at a position coinciding with the central axis in the longitudinal direction of rectangular waveguide 11, where the electric field intensity is at maximum strength, dielectric plate 13 exercises a large effect upon the propagation speed of radio waves, but the electric field strength progressively weakens with distance from the longitudinal central axis of the rectangular waveguide 11 and proximity to either of the short sides, and consequently, as the position of insertion of dielectric plate 13 shifts away from the central axis of rectangular waveguide 11 and approaches either of the short sides, the effect upon the propagation speed of radio waves decreases. Accordingly, the degree of phase shifting can be varied by changing the position of insertion of dielectric plate 13. Support fittings 14 1 and 14 2 are maintained parallel to each other, their relative spacing (spacing in relation to the longitudinal direction of rectangular waveguide 11) being selected as λ g /4 (λ g is the wavelength within the waveguide corresponding to the employed frequency), whereby reflected waves arising at the support fitting 14 1 closer to the radio-wave-incident portion and reflected waves arising at support fitting 14 2 which travel back to the position of support fitting 14 1 are of mutually reversed phase and cancel each other, thereby enabling an improved reflection characteristic. A line stretcher used in the prior art must regulate the line length according to the required degree of phase shifting, and therefore, when the required degree of phase shifting is great, not only is a large-scale mechanical structure required, but a relatively time-consuming and labor-intensive adjustment is required to accurately match the line length with the degree of phase shift. The waveguide-type phase shifting device shown in Fig. 1 involves the drawbacks that a long dielectric plate 13 is required when a large amount of phase shift is called for, resulting in a phase shifting device of large overall size, and that a great deal of time and effort is required to adjust the insertion point of the dielectric plate 13 to accurately match the amount of phase shift with the required value. Fig. 2 shows a power supply circuit configured using phase shifting devices of the prior art for controlling the driving phase of subarray antennas and for varying the direction of maximum radiation as well as the side lobe characteristic of an array antenna. This power supply circuit is configured from subarray antennas 16 1 -16 4 each composed of a plurality of element antennas, phase shifting devices 17 1 -17 3 of the above-described prior art, transmission lines 18 1 -18 3 having a degree of phase shift that serves as a standard, and two-branch circuits 19 1 -19 3 . A phase shifting device of the prior art not only entails the same drawbacks as the above-described line stretcher, but when used as shown in Fig. 2, in which a power supply circuit is configured for varying the direction of maximum radiation or side lobe characteristic of an array antenna, further entails the drawback of complex structure of the power supply circuit due to the need for transmission lines 18 1 -18 3 having an amount of phase shift that serves as a standard and two-branch circuits 19 1 -19 3 for phase shifting devices 17 1 -17 3 , respectively. DISCLOSURE OF THE INVENTION The object of the present invention is to provide a phase shifting device which has a simple and compact structure, and which allows easy adjustment at the time of manufacture and easy handling during use. GB-A-2 005 083 discloses a phase shifter for use at microwave frequencies comprising a first antenna arranged to emit circularly polarized waves from the microwave source and the second antenna for receiving the waves, the antennae being mounted for relative axial rotation. If signals must be transmitted efficiently from the first to the second antenna, then the size of the antenna has to be increased to an extent of few wavelengths to decrease the amount of leakage, resulting in failure provide a compact phase shifting device. DISCLOSURE OF THE INVENTION The object of the present invention is to provide a phase shifting device which has a simple and compact structure, and which allows easy adjustment at the time of manufacture and easy handling during use. The present invention provides a phase shifting device with the features of claim 1. The two-partition output of the two-partition circuit, which distributes the input power in two portions having a mutual phase difference of substantially 90°, is applied to the first microwave antenna means, which generates from its front surface a circularly polarized wave. This circularly polarized wave couples at the second antenna means, and the two outputs of the second antenna means arising from this coupled circularly polarized wave are combined at the combining circuit. In the following the first and second microwave antenna means are designated as first and second drives, respectively. When the first and second drivers are rotated relative to each other around the axis that joins the centers of each of the first and second drivers, the phase difference between the input of the two-partition circuit that applies driver power to the first driver and the combined output of the second driver varies in accordance with this rotation. This phase shifting device allows variation of the degree of phase shift over a range of from 0° to 360°, and moreover, has a simple and compact structure that enables extremely easy adjustment at the time of manufacture and service during use. In addition, the relation between relative angle of rotation of the two sealed cases and the degree of phase shift is proportional, and therefore, the degree of phase shift can be read directly if graduations of degrees are added around the circumference of one of the sealed cases and a pointer is added at a point on the circumference of the other sealed case. Furthermore, the present invention provides a phase shifting device with the features of claim 7. When the first driver is driven, linearly polarized waves are generated from the front surface of the driver, and the two orthogonal components of this linearly polarized wave are coupled at the second driver. When the first and second drivers are rotated relative to each other around the axis joining the centers of each of the first and second drivers, the phase difference between the input applied to the first driver and either of the two outputs of the combining circuit to which the two outputs of the second driver are applied varies according to the angle of rotation; and in addition, the phase difference between the two outputs of the combining circuit to which the two outputs of the second driver are applied also varies according to the relative angle of rotation between the first and second drivers. This phase shifting device allows variation of the degree of phase shift over a range of from 0° to 360°, and moreover, has a simple and compact structure and enables extremely easy adjustment at the time of manufacture and handling during use. In addition, the relation between relative angle of rotation of the two sealed cases and the degree of phase shift is proportional, and therefore, the degree of phase shift can be read directly if graduations of degrees are added around the circumference of one of the sealed cases and a pointer is added at a point on the circumference of the other sealed case. This phase shifting device is well-suited for control of the direction of maximum radiation or side lobe characteristics of an array antenna because it both allows variation of the phase difference between the input and either one of the two outputs of the combining circuit according to the relative angle of rotation between the sealed cases, and moreover, allows variation of the phase difference between the two outputs according to the relative angle of rotation between the sealed cases. BRIEF DESCRIPTION OF THE DRAWING Figs. 1A - 1D show an example of a waveguide-type phase shifting device of the prior art; Fig. 2 shows an example of the use of the prior-art phase shifting device shown in Fig. 1; Fig. 3 is a side view of a phase shifting device according to an embodiment of the present invention; Fig. 4 is a side view of a coupler 21 shown in Fig. 3; Fig. 5 is a plan view of coupler shown in Fig. 3; Fig. 6 is a sectional view taken at line C-C of Fig. 4; Fig. 7 is a sectional view taken at line D-D of Fig. 5; Fig. 8 illustrates the operation of the phase shifting device according to the embodiment of Fig. 3; Fig. 9 shows the return loss characteristic in coaxial connection plug 21 31 of the phase shifting device of the embodiment shown in Fig. 3; Fig. 10 shows the transmission loss characteristic between coaxial connection plugs 21 31 and 21 34 of the phase shifting device of the embodiment shown in Fig. 3; Fig. 11 shows the degree of phase shift in relation to the relative angle of rotation  between sealed cases 21 1 and 21 2 in the phase shifting device of the embodiment shown in Fig. 3; Fig. 12 shows the transmission loss characteristic between input terminal 22 1 and output terminal 24 1 in the phase shifting device of the embodiment shown in Fig. 3; Fig. 13 is a sectional view showing the principal parts of a phase shifting device according to another embodiment of the present invention; Fig. 14 is a sectional view showing the principal parts of a phase shifting device according to another embodiment of the present invention; Fig. 15 is a block diagram showing a phase shifting device according to the embodiment of Fig. 13; Fig. 16 is a side view of a phase shifting device according to another embodiment of the present invention; Fig. 17 is a side view of coupler 31 within Fig. 16; Fig. 18 is a plan view of coupler 31 within Fig. 16; Fig. 19 is a sectional view taken at line G-G of Fig. 17; Fig. 20 is a sectional view taken at line H-H of Fig. 17; Fig. 21 is a sectional view taken at line I-I of Fig. 18; Figs. 22A and 22B show the electric field level produced by drive element 31 and the outputs E 1 and E 2 of 90° 3-dB hybrid circuit 32 in the phase shifting device of Fig. 16; Fig. 23 shows the reflection characteristic in coaxial connection plug 31 31 of the phase shifting device of Fig. 16; Fig. 24 shows the transmission loss characteristic between coaxial connection plugs 31 31 and 31 33 of the phase shifting device of Fig. 16; Fig. 25 shows the degree of phase shift in relation to the relative angle of rotation  between sealed cases 31 1 and 31 2 in the phase shifting device of Fig. 16; Fig. 26 shows the observation results of the transmission loss between coaxial connection plug 31 31 of coupler 31 and the output terminal 32 3 of 90° 3-dB hybrid circuit 32, and between coaxial connection plug 31 31 of coupler 31 and output terminal 32 4 of 90° 3-dB hybrid circuit 32; Fig. 27 shows an example of the use of both outputs E 2 and E 1 of output terminals 32 3 and 32 4 of 90° 3-dB hybrid circuit 32; and Fig. 28 is a sectional view showing the principal parts of the phase shifting device according to another embodiment of the present invention. DETAILED DESCRIPTION OF THE PREFFERED EMBODIMENTS Fig. 3 is a side view of the phase shifting device according to an embodiment of the present invention. The phase shifting device according to this embodiment is constructed from coupler 21 which includes cylindrical sealed cases 21 1 and 21 2 which are each closed at one end, and terminals 21 31 -21 34 made up of coaxial connection plugs; 90° 3-dB hybrid circuit 22 which is composed of a directional coupler including input terminal 22 1 , isolation terminal 22 2 , and output terminals 22 3 and 22 4 ; reflectionless terminator 23 connected to isolation terminal 22 2 ; 90° 3-dB hybrid circuit 24 which is composed of a directional coupler including output terminal 24 1 , isolation terminal 24 2 , and input terminals 24 3 and 24 4 ; reflectionless terminator 25 connected to isolation terminal 24 2 ; coaxial cable 26 1 connecting output terminal 22 4 and terminal 21 31 ; coaxial cable 26 2 connecting output terminal 22 3 and terminal 21 32 ; coaxial cable 27 1 connecting terminal 21 34 and input terminal 24 3 ; and coaxial cable 27 2 connecting terminal 21 33 and input terminal 24 4 . Here, the output power of each of output terminals 22 3 and 22 4 is 1/2 of the input power to input terminal 22 1 , and, for example, the phase of the output of output terminal 22 3 is substantially 90° delayed with respect to the phase of the output of output terminal 22 4 . Furthermore, inputs of equivalent magnitude are applied to each of input terminals 24 3 and 24 4 , the input to input terminal 24 3 being, for example, substantially 90° advanced with respect to the phase of input to input terminal 24 4 . The lengths of each of coaxial cables 26 1 and 26 2 are adjusted such that the phase of the output of output terminal 22 3 of 90° 3-dB hybrid circuit 22 is delayed substantially 90° with respect to the phase of the output of output terminal 22 4 , and moreover, such that both outputs are applied to input coaxial connection plugs 21 31 and 21 32 of coupler 21 while being kept in a mutually equivalent relation. The lengths of coaxial cables 27 1 and 27 2 are each adjusted such that the outputs from output coaxial connection plugs 21 33 and 21 34 of coupler 21 are applied to the input terminals 24 3 and 24 4 of 90° 3-dB hybrid circuit 24 with the phase difference and the equal amplitude relation between the outputs maintained unchanged. Fig. 4 shows a side view of coupler 21 of Fig. 3, Fig 5 shows a plan view of the sealed case 21 1 side of coupler 21 as seen from the bottom side, Fig. 6 is an enlarged sectional view taken along line C-C of Fig. 4, and Fig. 7 is an enlarged sectional view taken along line D-D of Fig. 5. As can be clearly seen from Fig. 7, stepped portions are formed in the side walls at the end portions of the openings of both of sealed cases 21 1 and 21 2 , the end portion of the opening of sealed case 21 2 fitting inside the end portion of the opening of sealed case 21 1 , the two sealed cases 21 1 and 21 2 being coupled together as a unit, and sealed cases 21 1 and 21 2 being constructed so as to allow rotation relative to each other around their cylindrical axis. Sealed cases 21 1 and 21 2 are manufactured by first machining a metal block or press-forming a metal sheet in the prescribed shape; by forming a suitable synthetic resin in a preliminary form of the prescribed shape and subsequently applying a metal film to the surface by a electroless plating or vacuum evaporation process; or by multilayer stacking of dielectric plates having an appropriate dielectric constant and forming as a cylinder with one closed end that shields electromagnetic energy in accordance with Snell's law. Coaxial connection plugs 21 31 and 21 32 each include internal conductors 21 41 and 21 42 . A round insulation plate 21 51 composed of a material having excellent highfrequency characteristics and a thickness less than the transmission wavelength such as an organic material such as polyethylene or fluorinated ethylene or an inorganic material such as a ceramic is adhered to the inner surface of the bottom wall of sealed case 21 1 using a suitable adhesive. Metal plate 21 61 is secured to the surface of round insulation plate 21 51 using a suitable adhesive, and in addition, portions of the periphery of metal plate 21 61 are connected to internal conductors 21 41 and 21 42 of coaxial connection plugs 21 31 and 21 32 , and the first driver composed of a microstrip antenna is formed with sealed case 21 1 on the opposite side of round insulation plate 21 51 as the ground conductor, metal plate 21 61 as the first driver element, and the contact points between metal plate 21 61 and interior conductors 21 41 and 21 42 of coaxial connection plugs 21 31 and 21 32 as the driver points. As can be seen from Fig. 6, a right angle is formed by the straight line joining the center of metal plate 21 61 and the driving point constituted by the contact point between metal plate 21 61 and the interior conductor 21 41 of coaxial connection plug 21 31 and the straight line joining the center of metal plate 21 61 and the driving point constituted by the contact point between metal plate 21 61 and the internal conductor 21 42 of coaxial connection plug 21 31 . In addition, the exterior conductors of each of coaxial connection plugs 21 31 and 21 32 are electrically connected to sealed case 21 1 , and as shown in Fig. 7, a portion of sealed case 21 1 has been removed around the periphery of internal conductor 21 41 to form a gap such that internal conductor 21 41 makes no mechanical contact with sealed case 21 1 , and a gap is similarly formed in sealed case 21 1 at the portion around the periphery of internal conductor 21 42 . As the method for securing metal plate 21 61 to the surface of round insulation plate 21 51 , a setscrew in the center of metal plate 21 61 (the portion wherein electric field intensity is 0) may also be used to secure the plate to round insulation plate 21 51 rather than an adhesive, or metal plate 21 61 may be screwed at its center to the bottom wall of sealed case 21 1 through round insulation plate 21 51 . Instead of using metal plate 21 61 , the driver element of the microstrip antenna making up the first driver may be formed by applying a metal film to the surface of round insulation plate 21 51 by a process such as vacuum evaporation and finishing to the required outline shape by a process such as etching. Whether the driver element is formed from a metal plate or metal film, its outline shape may instead of a circle be formed as a square, electrical connection with the internal conductors 21 41 and 21 42 of coaxial connection plugs 21 31 and 21 32 being effected in the corner portions of the square. Next will be explained the dimensions of driver element 21 61 for a case in which driver element 21 61 is driven in basic mode. When the outline shape of driver element 21 61 is formed as a circle as shown in Fig. 6, if the specific inductive capacity of round insulation plate 21 51 is ε r and the design frequency is f o , the radius of driver element 21 61 is preferably selected as substantially 1.841 C/(2π · f o · ε r 1/2 ), and when the outline shape of driver element 21 61 is formed as a square, the length of one side of driver element 21 61 is preferably selected as substantially C/(2f o · ε r 1/2 )(C being the speed of light). Next, in Fig. 7, 21 43 and 21 44 are the internal conductors of coaxial connection plugs 21 33 and 21 34 , respectively, 21 52 is a round insulation plate, and 21 62 is the second driver element; the material, method of fabrication, and mutual electrical and mechanical relation of each of these parts being identical to the material, method of fabrication, and mutual electrical and mechanical relation of each of respective coaxial connection plugs 21 31 and 21 32 , internal conductors 21 41 and 21 42 , round insulation plate 21 51 , and first driver element 21 61 on the side of sealed case 21 1 . Fig. 8 illustrates the operation of this phase shifting device. The input power E applied to the input terminal of 90° 3-dB hybrid circuit 22 is distributed into two substantially equal portions and outputted from output terminals 22 3 and 22 4 , the phase of output E 1 of output terminal 22 3 being delayed substantially 90° with respect to the phase of output E 2 of output terminal 22 4 . Output is outputted to isolation terminal 22 2 according to the difference between the impedance viewing the load side from output terminal 22 3 and the impedance viewing the load side from output terminal 22 4 , and this output is absorbed by reflectionless terminator 23; however, in a state in which output terminal 22 3 and the load are matched and output terminal 22 4 and the load are matched, the output outputted to isolation terminal 22 2 becomes extremely small, and as a result, the permissible power of reflectionless terminator 23 may be extremely low. Output E 1 of output terminal 22 3 of 90° 3-dB hybrid circuit 22 is applied by way of coaxial cable 26 2 to the driver point at which internal conductor 21 42 of coaxial connection plug 21 32 of coupler 21 connects to first driver element 21 61 . The instantaneous value of the electric field propagated from this driving point in the Z direction (the X-axis and Y-axis are determined as shown in Fig. 6, and the direction perpendicular to the X- and Y-axis is the Z direction) is E X . Output E 2 of output terminal 22 4 of 90° 3-dB hybrid circuit 22 is applied by way of coaxial cable 26 1 to the driver point at which internal conductor 21 41 of coaxial connection plug 21 31 of coupler 21 is connected to first driver element 21 61 , and the instantaneous value of the electric field propagated in the Z direction from this driver point is E Y . The instantaneous values E X and E Y of the electric field are: E 1 = - j E 2 ≡ E X E 2 = E 2 ≡ E Y The electric field propagated in the Z direction couples at the second driver element 21 62 , which together with round insulation plate 21 52 and the bottom wall of the second sealed case 21 2 forms the second driver; but if the angle formed between the straight line joining the center of first driver element 21 61 with the driver point at which internal conductor 21 41 of coaxial connection plug 21 31 of coupler 21 connects to first driver element 21 61 and the straight line joining the center of second driver element 21 62 with the driver point at which internal conductor 21 44 of coaxial connection plug 21 34 of coupler 21 connects to second driver element 21 62 is represented by , the output of coaxial connection plug 21 34 is represented by E 3 , and the output of coaxial connection plug 21 33 is represented by E 4 , the outputs E 3 and E 4 are: E 3 = E Y cos  - E X sin  E 4 = E Y sin  - E X cos  The output E 3 of coaxial connection plug 21 34 is applied through input terminal 24 3 to 90° 3-dB hybrid circuit 24, and output E 4 of coaxial connection plug 21 33 is applied through input terminal 24 4 to 90° 3-dB hybrid circuit 24. If the output of isolation terminal 24 2 of 90° 3-dB hybrid circuit 24 is represented by E 5 , and the output of output terminal 24 1 is represented by E 6 , outputs E 5 and E 6 are respectively: In other words, in this phase shifting device, the phase of the output of output terminal 24 1 of 90° 3-dB hybrid circuit 24 can be shifted by  with respect to the phase of the input to input terminal 22 1 of 90° 3-dB hybrid circuit 22 by simply rotating the sealed cases 21 1 and 21 2 with respect to each other by the angle . Fig. 9 shows the reflection characteristics at coaxial connection plug 21 31 for a case in which the inside diameter of the cylinders having one closed end that form sealed cases 21 1 and 21 2 of coupler 21 are selected as 0.285 λ o (λ o being the free space wavelength corresponding to the designed frequency f o ), the distance between the bottom wall of sealed case 21 1 and the bottom wall of sealed case 21 2 is set at 0.089 λ o , the specific inductive capacity of each of round insulation plates 21 51 and 21 52 is set at 10, the dielectric dissipation factor of each of round insulation plates 21 51 and 21 52 is set at 0.0055, the thickness of round insulation plates 21 51 and 21 52 is set at 0.023 λ o , and each of the first and second driver elements 21 61 and 21 62 are formed in a circular shape of diameter 0.21 λ o . Here, the axis of abscissas shows the relative frequency with respect to the design frequency f o , and the axis of ordinates shows the amount of return loss (dB). Fig. 10 shows the transmission characteristics between coaxial connection-plugs 21 31 and 21 34 for a case in which each of the dimensions of coupler 21 is the same as for Fig. 9, the axis of abscissas being equivalent to Fig. 9 and the axis of ordinates showing the amount of transmission loss (dB). As can be seen from Fig. 9 and Fig. 10, coupler 21 exhibits excellent reflection characteristics and transmission characteristics over a broad band. Furthermore, the thickness of round insulation plates 21 51 and 21 52 of coupler 21 can be selected according to the transmission frequency band, and the bandwidth can be broadened by increasing the thickness from the selected thickness. Fig. 11 shows observation results of changes in phase of the output from output terminal 24 1 of 90° 3-dB hybrid circuit 24 with respect to the phase of the input to input terminal 22 1 of 90° 3-dB hybrid circuit 22, the axis of abscissas showing the frequencies relative to the design frequency f o and the axis of ordinates showing the amount of phase shift ( · ) with respect to the angle of rotation , each of the dimensions of coupler 21 being selected in the same way as explained for Fig. 9, the phase shifting device being configured with 90° 3-dB hybrid circuit 22 connected to coupler 21 by way of coaxial cables 26 1 and 26 2 and 90° 3-dB hybrid circuit 24 connected to coupler 21 by way of coaxial cables 27 1 and 27 2 as shown in Fig. 3, and  representing the angle of rotation by which the sealed cases 21 1 and 21 2 forming coupler 21 have been rotated relative to each other around their common cylindrical axis. As is clear from Fig. 11, regardless of the transmission frequency, the amount of phase shift always coincides with the angle of rotation  of sealed cases 21 1 and 21 2 making up coupler 21 with respect to each other around their common cylindrical axis. Fig. 12 shows the observation results of the amount of transmission loss between input terminal 22 1 of 90° 3-dB hybrid circuit 22 and output terminal 24 1 of 90° 3-dB hybrid circuit 24 under the same conditions as for the observations in which the results of Fig. 11 were obtained, the axis of abscissas being the same as for Fig. 11 and the axis of ordinates showing the amount of transmission loss (dB). As is clear from Fig. 12, compared to the amount of transmission loss shown in Fig. 10 for coupler 21 alone, the addition of 90° 3-dB hybrid circuits 22 and 24, coaxial cables 26 1 and 26 2 , as well as 27 1 and 27 2 increases the transmission loss, but the degree of change in the frequency characteristic of the amount of transmission loss is as for coupler 21 alone. The foregoing explanation relates to a case in which sealed cases 21 1 and 21 2 of coupler 21 are formed from metal; the first driver installed within coupler 21 is formed from a microstrip antenna composed of sealed case 21 1 , which is the ground conductor, round insulation plate 21 51 , and driver element 21 61 ; and the second driver is formed from a microstrip antenna composed of sealed case 21 2 , which is the ground conductor, round insulation plate 21 52 , and driver element 21 62 . However, in a case in which sealed cases 21 1 and 21 2 are formed by applying a metal film to the outer surface of a body composed of a suitable synthetic resin, the metal film applied to the outer surface of the body of sealed case 21 1 can serve as the ground conductor, the body of sealed case 21 1 can serve as round insulation plate 21 51 , and the first driver composed of a microstrip antenna can therefore be formed by attaching driver element 21 61 to the inner surface of the bottom wall of sealed case 21 1 , and the second driver composed of a microstrip antenna can similarly be formed by attaching driver element 21 62 to the inner surface of the bottom wall of sealed case 21 2 . In a case in which sealed cases 21 1 and 21 2 are formed by multilayer stacking of dielectric plates, the first driver composed of a microstrip antenna can be formed by attaching driver element 21 61 to the inner surface of the bottom wall of sealed case 21 1 and attaching a ground conductor to the outer surface of the bottom wall; and the second driver composed of a microstrip antenna can be formed by attaching driver element 21 62 to the inner surface of the bottom wall of sealed case 21 2 and attaching a ground conductor to the outer surface of the bottom wall. Instead of forming the driver as a microstrip antenna, the driver can also be formed from a slot antenna formed by providing a cross-shaped slot 21 71 in the central portion of metal plate or metal film 21 61 forming the driver element as shown in Fig. 13, which is a sectional view taken along the same line as Fig. 6. Other construction and reference numerals in Fig. 13 are as for Fig. 6. In Fig. 14, which is a sectional view taken along the same line as Fig. 7, 21 71 is cross-shaped slot, and 21 72 , like 21 71 , is a cross-shaped slot provided in the central portion of the metal plate or metal film 21 62 forming the driver element, the other construction and reference numerals being the same as for Fig. 7. In this embodiment as well, if sealed cases 21 1 and 21 2 are formed by applying a metal film to the outer surface of a base body composed of a suitable synthetic resin, the metal film applied to the outer surface of the base body can serve as the ground conductor and the base body itself can be used as round insulation plates 21 51 and 21 52 of Fig. 14; and if sealed cases 21 1 and 21 2 are formed by multilayer stacking of dielectric plates, the first and second drivers composed of slot antennas can be formed by attaching driver elements 21 61 and 21 62 composed of metal films or metal plates provided with cross-shaped slots to the inner surface of each of the bottom walls of sealed cases 21 1 and 21 2 , and providing a ground conductor on the outer surface of each bottom wall. Further, probes may be connected to internal conductors 21 41 and 21 42 of coaxial connection plugs 21 31 and 21 32 shown in Fig. 6 or Fig. 13, these probes being formed such that the longitudinal direction of both probes is parallel to metal plate or metal film 21 61 , the portions of both probes extending in the longitudinal direction intersect at the center point of metal plate or metal film 21 61 , and this intersecting angle is a right angle. A driver composed of probes similar to the sealed case 21 1 side may then be provided on the sealed case 21 2 side. Whichever of the above constructions is adopted for the drivers installed in each of sealed cases 21 1 and 21 2 , the amount of phase shift between the input and output is determined by the angle of relative rotation between sealed cases 21 1 and 21 2 , and in order for sealed cases 21 1 and 21 2 to rotate with respect to each other, any of the following constructions may be adopted: coaxial cables 26 1 , 26 2 , 27 1 , and 27 2 are formed as flexible cables; the sealed case 21 2 side is fixed and sealed case 21 1 , coaxial cables 26 1 and 26 2 , and 90° 3-dB hybrid circuit 22 rotate as a unit; the sealed case 21 1 side is fixed and sealed case 21 2 , coaxial cables 27 1 and 27 2 , and 90° 3-dB hybrid circuit 24 rotate as a unit; or sealed case 21 1 , coaxial cables 26 1 and 26 2 , and 90° 3-dB hybrid circuit 22 are formed as one rotatable unit and sealed case 21 2 , coaxial cables 27 1 and 27 2 and 90° 3-dB hybrid circuit 24 are formed as another rotatable unit. The foregoing explanations relate to cases in which sealed cases 21 1 and 21 2 are each formed as a cylinder with one closed end, but either of sealed case 21 1 or 21 2 may be formed as a cylindrical case with one closed end in which is installed either the first or second driver, and the other side, i.e., sealed case 21 2 or 21 1 , may be formed as a disk-shaped cover with the second or first driver attached to the inner surface, this cover being formed to rotatably fit with the end portion of the opening of the closed cylinder body. As is clear from the foregoing explanation, in this phase shifting device, the side of coaxial cables 26 1 and 26 2 and 90° 3-dB hybrid circuit 22 and the side of coaxial cables 27 1 and 27 2 and 90° 3-dB hybrid circuit 24 are formed symmetrically to each other across the plane cutting through the midsection of the coupler 21 at F-F in Fig. 15; and consequently, precisely identical operation can be achieved as explained in conjunction with Fig. 8 even if terminal 24 1 of 90° 3-dB hybrid circuit 24 is made the input terminal and terminal 22 1 of 90° 3-dB hybrid circuit 22 is made the output terminal. The other reference numerals of Fig. 15 are equivalent to those of Fig. 3. The foregoing explanation relates to an example in which 90° 3-dB hybrid circuits 22 and 24 and reflectionless terminators 23 and 25 have been used for the two-partition circuit of the input and the combining circuit of the output, but for a case in which the transmission frequency bandwidth is relatively narrow, the input- and output-side 90° 3-dB hybrid circuits 22 and 24 and reflectionless terminators 23 and 25 can be replaced by two-branch terminal circuits, with one of coaxial cables 26 1 and 26 2 connecting coaxial connection plugs 21 31 and 21 32 of coupler 21 with the two output terminals of the two-branch terminal circuit on the input side, for example, coaxial cable 26 2 , being formed exactly one transmission quarter-wavelength longer than coaxial cable 26 1 , and one of coaxial cables 27 1 and 27 2 connecting coaxial connection plugs 21 33 and 21 34 of coupler 21 to the two input terminals of the two-branch terminal circuit on the output side, for example coaxial cable 27 1 , being formed exactly one transmission quarter-wavelength longer than coaxial cable 27 2 . In this embodiment, although the lengths of coaxial cables 26 1 and 26 2 must be made to differ by exactly one transmission quarter-wavelength, and the lengths of coaxial cables 27 1 and 27 2 similarly must be made to differ by exactly one transmission quarter-wavelength, the difference in lengths for both sets of cables is fixed at one transmission quarter-wavelength, and fabrication is therefore relatively easy. If the difference in lengths between coaxial cables 26 1 and 26 2 and the difference in lengths between coaxial cables 27 1 and 27 2 is selected to be one quarter-wavelength of the center frequency of the transmission band, the difference in length between the coaxial cables will not precisely match the quarter-wavelength for frequencies outside the center frequency, but since this embodiment is intended for applications in which the transmission frequency band is relatively narrow, any operational error arising due to variance from the quarter-wavelength is minute and presents no practical problem. The foregoing explanation relates to a case in which the phase shifting device of the present invention is constructed by assembling three-dimensional constituent elements, but the entire structure can be miniaturized by forming 90° 3-dB hybrid circuits 22 and 24 and reflectionless terminators 23 and 25 on a printed circuit board using a printed wiring method, and forming coaxial cables 26 1 , 26 2 , 27 1 and 27 2 as microstrip wiring. The entire structure may also be made extremely compact and concise by providing dielectric layers on the outer surfaces of sealed cases 21 1 and 21 2 of coupler 21, and then forming by a printed wiring method on the dielectric layer provided on the outer surface of sealed case 21 1 microstrip wiring that takes the place of 90° 3-dB hybrid circuit 22, reflectionless terminator 23 and coaxial cables 26 1 and 26 2 , and forming by a printed wiring method on the dielectric layer provided on the outer surface of sealed case 21 2 microstrip wiring that takes the place of 90° 3-dB hybrid circuit 24, reflectionless terminator 25 and coaxial cables 27 1 and 27 2 . In a case in which two-branch terminal circuits are used in place of 90° 3-dB hybrid circuits 22 and 24 and reflectionless terminators 23 and 25, the entire structure can be miniaturized through formation on printed circuit boards or on dielectric layers provided on the outer surfaces of each of sealed cases 21 1 and 21 2 . Fig. 16 presents a phase shifting device according to another embodiment of the present invention. The phase shifting device of this embodiment is constructed from a coupler 31 composed of cylindrical sealed cases 31 1 and 31 2 having one closed end, and terminals 31 31 -31 33 composed of, for example, coaxial connection plugs; 90° 3-dB hybrid circuit 32 composed of, for example, a quarter-wave coupled line-type directional coupler including input terminal 32 1 , isolation terminal 32 2 , and output terminals 32 3 and 32 4 ; coaxial cable 33 1 connecting terminal 31 33 and input terminal 32 1 ; and coaxial cable 33 2 connecting terminal 31 32 and isolation terminal 32 2 . Here, considering a signal inputted to input terminal 32 1 , the phase of the output from output terminal 32 3 is delayed substantially 90° with respect to the phase of the output of output terminal 32 4 . In addition, the length of each of coaxial cable 33 1 and 33 2 is adjusted such that the phase difference between inputs at the input end of each of coaxial cables 33 1 and 33 2 is maintained unchanged when outputted from the output end of each of coaxial cables 33 1 and 33 2 . Fig. 17 shows a side view of coupler 31 of Fig. 16, and Fig. 18 shows a plan view of the sealed case 31 1 side of coupler 31 as seen from the side of the bottom wall, the reference numerals used in Figs. 17 and 18 being the same as those used in Fig. 16. Fig. 19 is an enlarged sectional view taken at line G-G of Fig. 16, Fig. 20 is an enlarged sectional view taken at line H-H of Fig. 17, and Fig. 21 is an enlarged sectional view taken at line I-I of Fig. 18. As can be clearly seen from Fig. 21, stepped portions are formed in the side walls at the end portions of the openings of each of sealed cases 31 1 and 31 2 , the end portion of the opening of one of the sealed cases, in this case sealed case 31 2 , fitting inside the end portion of the opening of the other sealed case 31 1 , the two sealed cases 31 1 and 31 2 being mechanically and electrically coupled, and sealed cases 31 1 and 31 2 able to rotate relative to each other around their cylindrical axis. Sealed cases 31 1 and 31 2 are manufactured by machining a metal block or press-forming a metal sheet in the prescribed shape; by forming a suitable synthetic resin in a preliminary form of the prescribed shape and subsequently applying a metal film to the surface by an electroless plating or vacuum evaporation process; or by multilayer stacking of dielectric plates having an appropriate dielectric constant and forming as a cylinder with one closed end that shields electromagnetic energy in accordance with Snell's law. Coaxial connection plugs 31 31 to 31 33 have internal conductors 31 41 to 31 43 . Round insulation plates 31 51 and 31 52 are composed of a material having excellent highfrequency characteristics such as an organic material such as polyethylene or fluorinated ethylene or an inorganic material such as a ceramic and have a thickness less than the transmission wavelength, round insulation plate 31 51 being secured to the inner surface of the bottom wall of sealed case 31 1 , round insulation plate 31 52 being secured to the inner surface of the bottom wall of sealed case 31 2 , and each being secured using a suitable adhesive. Metal plate 31 61 is secured to the surface of round insulation plate 31 51 using a suitable adhesive, and in addition, a portion of its periphery is electrically connected to internal conductor 31 41 of coaxial connection plug 31 31 . First driver composed of a microstrip antenna is formed with sealed case 31 1 on the opposing side of insulation plate 31 51 serving as ground conductor, metal plate 31 61 serving as first driver element, and the connection point between internal conductor 31 41 of coaxial connection plug 31 31 and metal plate 31 61 serving as the driver point. Metal plate 31 62 is secured to the surface of round insulation plate 31 52 using a suitable adhesive, and in addition, portions of its periphery are electrically connected to internal conductors 31 42 and 31 43 of coaxial connection plugs 31 32 and 31 33 . Second driver composed of a microstrip antenna is formed with sealed case 31 2 on the opposing side of insulation plate 31 52 serving as ground conductor, metal plate 31 62 serving as second driver element, and the connection points between metal plate 31 62 and internal conductors 31 42 and 31 43 of coaxial connection plugs 31 32 and 31 33 serving as the driver points. As can be seen from Fig. 20, a right angle is formed by the intersection of the straight line joining the center of metal plate 31 62 with the driving point composed of the connection point between internal conductor 31 42 of coaxial connection plug 31 32 and metal plate 31 62 , and the straight line joining the center of metal plate 31 62 with the driving point formed by the connection point between internal conductor 31 43 of coaxial connection plug 31 33 and metal plate 31 62 . In addition, the spacing between first driver element 31 61 and second driver 31 62 is set at less than the transmission wavelength. The external conductor of coaxial connection plug 31 31 is electrically connected to sealed case 31 1 , and the external conductors of each of coaxial connection plugs 31 32 and 31 33 are connected to sealed case 31 2 ; but, as shown in Fig. 21, a gap is provided by removing a portion of the sealed case 31 1 in the vicinity of internal conductor 31 41 and a gap is provided by removing a portion of sealed case 31 2 in the vicinity of internal conductor 31 43 such that internal conductors 31 41 and 31 43 of coaxial connection plugs 31 31 and 31 33 do not mechanically contact sealed cases 31 1 and 31 2 . Although not shown in Fig 21, a gap is similarly provided in sealed case 31 2 in the vicinity of internal conductor 31 42 of coaxial connection plug 31 21 . Rather than using an adhesive as a means of securing metal plate 31 61 or 31 62 to the surface of round insulation plate 31 51 or 31 52 , the center of metal plate 31 61 or 31 62 (the portion where electric field intensity is 0) may be secured to round insulation plate 31 51 or 31 52 using a setscrew, or alternatively the center of metal plate 31 61 or 31 62 may be screwed to the bottom wall of sealed case 31 1 or 31 2 through round insulation plate 31 51 or 31 52 . Rather than forming the driver element of the microstrip antenna constituting the first or second driver from metal plate 31 61 or 31 62 , the driver element may be formed by applying a metal film to the surface of round insulation plate 31 51 or 31 52 using a vacuum evaporation process and finishing to the required outline shape by a process such as etching. Whether the driver element is formed from a metal plate or metal film, its outline shape may be formed as a square instead of the circle shown in Fig. 19 and Fig. 20. Regarding the dimensions of driver elements 31 61 and 31 62 for cases in which driver elements 31 61 and 31 62 are driven in basic mode, if the outline shape of these elements is formed as a circle, the specific inductive capacity of round insulation plates 31 51 and 31 52 is ε r and the design frequency is f o , the radius of driver elements 31 61 and 31 62 is preferably selected to be substantially 1.841 C/(2π · f o · ε r 1/2 ); and if the outline shape of driver elements 31 61 and 31 62 is formed as a square, the length of one side is preferably selected to be substantially C/(2f o · ε r 1/2 )(where C is the speed of light). Fig. 22 illustrates the operation of the present phase shifting device. The input applied to coaxial connection plug 31 31 of coupler 31 passes through internal conductor 31 41 and drives driver element 31 61 in the first driver. If the X-axis and Y-axis are established as shown in Fig. 19, Fig. 20, and Fig. 22A, electric field vector E produced by driver element 31 61 driven as described above can be divided between two orthogonal components E X and E Y as shown in Fig. 22A. These couple at driver element 31 62 of the second driver in the form: E = E X + E Y = E cos  + sin  As described hereinabove, this device is formed such that a right angle is formed by the intersection of the straight line joining the center of driver element 31 62 with the driving point constituted by the connection point between driver element 31 62 and internal conductor 31 42 of coaxial connection plug 31 32 and the straight line joining the center of driver element 31 62 with the driving point constituted by the connection point between driver element 31 62 and internal element 31 43 of coaxial connection plug 31 33 ; and therefore, when driver element 31 62 is driven in basic mode, coupling between coaxial connection plugs 31 32 and 31 33 becomes sparse and quadrature mode coupling is enabled. The quadrature mode coupling to driver element 31 62 causes component E X to be outputted from coaxial connection plug 31 32 and component E Y to be outputted from coaxial connection plug 31 33 , the two outputs being respectively inputted to terminals 32 1 and 32 2 of 90° 3-dB hybrid circuit 32 by way of coaxial cables 33 1 and 33 2 . If the output of output terminals 32 4 and 32 3 of 90° 3-dB hybrid circuit 32 are E 1 and E 2 , respectively: E 1 = E 2 sin  - j E 2 cos  = E 2 e -j{(π/2)-} E 2 = E 2 cos  - j E 2 sin  = E 2 e -jπ Although the phase of output E 1 shifts in a counterclockwise direction with increase in declination angle  of electric field vector E from the X-axis as shown in Fig. 22A, the amplitude remains unchanged. The phase of output E 2 also shifts in a clockwise direction with increase in declination angle  of electric field vector E from the X-axis, but again, the amplitude remains unchanged. Accordingly, if the declination angle  of electric field vector E from the X-axis is 45° or 225°, the amplitude as well as the phase of both outputs E 1 and E 2 are equal, but if the declination angle  increases from 45° or 225°, the phase of output E 1 shifts in the advance direction, and the phase of output E 2 shifts in the delay direction. If the declination angle  of electric field vector E from the X-axis further increases to 135° or 315°, the phase relation between outputs E 1 and E 2 is mutually reversed. Fig. 23 shows the reflection characteristic at coaxial connection plug 31 31 observed after removing coaxial cables 33 1 and 33 2 from coaxial connection plugs 31 32 and 31 33 , the inner diameter of the cylinders with one closed end forming sealed cases 31 1 and 31 2 of coupler 31 being selected as 0.285 λ o (λ o being the free-space wavelength corresponding to the design frequency f o ), the distance of opposition between the bottom wall of sealed case 31 1 and the bottom wall of sealed case 31 2 as 0.089 λ o , the specific inductive capacity of round insulation plates 31 51 and 31 52 as 10, the dielectric dissipation factor of each of round insulation plates 31 51 and 31 52 as 0.0055, the thickness of each of round insulation plates 31 51 and 31 52 as 0.023 λ o , the outline shape of first and second driver elements 31 61 and 31 62 being formed as a circle with a diameter of 0.21 λ o , the axis of abscissas in the figure being the relative frequency for the design frequency f o , and the axis of ordinates being the amount of return loss (dB). Fig. 24 shows the transmission characteristics between coaxial connection plugs 31 31 and 31 33 observed with coaxial cable 33 1 and 33 2 removed from coaxial connection plugs 31 32 and 31 33 and the selected values of the dimensions of each of the parts of coupler 31 being the same as described for Fig. 23, the horizontal axis of the figure being the same as for Fig. 23 and the vertical axis being the amount of transmission loss (dB). As is clear from Fig. 23 and Fig. 24, coupler 31 exhibits excellent reflection characteristics and transmission characteristics across a broad band. In addition, the thickness of round insulation plates 31 51 and 31 52 in coupler 31 can be selected according to the transmission frequency band, and the bandwidth can be broadened by increasing the selected thickness. Fig. 25 shows the results of observing the phase shift of output E 1 of coaxial connection plug 31 33 and output E 2 of coaxial connection plug 31 32 upon change of declination angle  to 90° or 135° with 45° as a standard. Here, the same values are selected for the dimensions of each part of coupler 31 as described for Fig. 23; the phase shifting device is configured as shown in Fig. 16, wherein 90° 3-dB hybrid circuit 32 is connected to coupler 31 by way of coaxial cables 33 1 and 33 2 ; and sealed cases 31 1 and 31 2 , which make up coupler 31, are rotated relative to each other around their common cylindrical axis. The angle of declination  is formed by the straight line joining the center of driver element 31 62 with the connection point between internal conductor 31 42 of coaxial connection plug 31 32 and driver element 31 62 in the second driver installed on the sealed case 31 2 side, i.e., the X-axis, and the electric field vector E arising due to drive of driver element 31 61 in the first driver installed on the sealed case 31 1 side. For both outputs E 1 and E 2 , the absolute value of the change in phase and the change in declination angle  match without any dependence on transmission frequency, and the code of phase of outputs E 1 and E 2 differ. Accordingly, the phase difference between outputs E 1 and E 2 with respect to change in declination angle  is 2. The axis of abscissas of Fig. 25 is the same as for-Fig. 23, and the axis of ordinates shows declination angle , i.e., the amount of phase shift (·) with respect to the relative angle of rotation  between sealed cases 31 1 and 31 2 . Fig. 26 shows the amount of transmission loss between coaxial connection plug 31 31 of coupler 31 and output terminal 32 3 of 90° 3-dB hybrid circuit 32, and between coaxial connection plug 31 31 of coupler 31 and output terminal 32 4 of 90° 3-dB hybrid circuit 32, the observation results being obtained under the same conditions as those for Fig. 25, the axis of abscissas being the same as Fig. 25 and the axis of ordinates showing the amount of transmission loss (dB). As is clear from Fig. 26, although the addition of 90° 3-dB hybrid circuit 32 and coaxial cables 33 1 and 33 2 results in increased transmission loss as compared with the amount of transmission loss for the case shown in Fig. 24 in which coupler 31 alone is used, the differences in the change of phase shift of outputs E 1 and E 2 with respect to the change in angle  or in amplitude of outputs E 1 and E 2 are virtually negligible. Fig. 27 shows an example which employs both outputs E 2 and E 1 of output terminals 32 3 and 32 4 of 90° 3-dB hybrid circuit 32. Reference numeral 34 indicates the phase shifting device shown in Fig. 16, 31 31 is the input terminal of coupler 31 in phase shifting device 34, 32 3 and 32 4 are the output terminals of 90° 3-dB hybrid circuit 32 in phase shifting device 34, and 35 1 and 35 2 are both subarray antennas composed of a plurality of element antennas. The locus joining the peaks of each of output vectors E X and E Y corresponding to the change in the relative angle of rotation  between sealed cases 31 1 and 31 2 which make up coupler 31 in phase shifting device 34 is an oval, but the angle at which output vectors E X and E Y begin to describe an oval both differs by 90° and is in the opposite direction of rotation, and consequently, the phase difference between outputs E 2 and E 1 can be freely selected. Accordingly, the direction of maximum radiation and the side-lobe characteristics of the array antenna composed of subarray antennas 35 1 and 35 2 can be controlled by supplying outputs E 2 and E 1 of output terminals 32 3 and 32 4 as the driving power of subarray antennas 35 1 and 35 2 . A comparison of Fig. 27 and Fig. 2 clearly shows that, while the structure of a power supply circuit using a phase shifting device of the prior art was complicated by the requirement for a transmission path having a standard amount of phase shift and a two-branch circuit for each phase shifter, the use of a phase shifting device of the present invention results in a power supply circuit of extremely simple construction because the phase shifting device of the present invention has an effect equivalent to the combination of a prior-art phase shifting device, a transmission path having a standard amount of phase shift, and a two-branch circuit. The foregoing description relates to an example in which sealed cases 31 1 and 31 2 of coupler 31 are formed from metal; the first driver installed in coupler 31 is formed from a microstrip antenna composed of sealed case 31 1 which serves as a ground conductor, round insulation plate 31 51 , and driver element 31 61 ; and the second driver is formed from a microstrip antenna composed of sealed case 31 2 which serves as a ground conductor, round insulation plate 31 52 , and driver element 31 62 . However, if sealed cases 31 1 and 31 2 are each formed by applying a metal film to the outer surface of a base body composed of a suitable synthetic resin, the metal film applied to the outer surface of the base body of sealed case 31 1 can be used as a ground conductor, the base body itself of sealed case 31 1 can be used as round insulation plate 31 51 , and as a result, the first driver composed of a microstrip antenna can be formed by attaching driver element 31 61 to the inner surface of the bottom wall of sealed case 31 1 , and the second driver composed of a microstrip antenna can be formed by attaching driver element 31 62 to the inner surface of the bottom wall of sealed case 31 2 . For a case in which sealed cases 31 1 and 31 2 are formed by multilayer stacking of dielectric plates, the first driver composed of a microstrip antenna can be formed by attaching driver element 31 61 to the inner surface and attaching a ground conductor to the outer surface of the bottom wall of sealed case 31 1 , and the second driver composed of a microstrip antenna can be formed by attaching driver element 31 62 to the inner surface and attaching a ground conductor to the outer surface of the bottom wall of sealed case 31 2 . As shown in Fig. 28 in a sectional view similar to that of Fig. 20, instead of being formed from a microstrip antenna, the second driver may be formed from a slot antenna achieved by providing a cross-shaped slot 31 7 in the center of metal plate or metal film 31 62 forming the driver element; and the first driver may be formed as a slot antenna composed of only a slot in the vertical direction or in the horizontal direction of the cross-shaped slot of Fig. 28. Other reference numerals and construction in Fig. 28 is as for Fig. 20. In this embodiment as well, if sealed cases 31 1 and 31 2 are formed by applying a metal film to the outer surface of a base body composed of a suitable synthetic resin, the metal film applied to the outer surface of the base body may be used as the ground conductor, and the base body itself can be used as round insulation plates 31 51 and 31 52 ; and if sealed cases 31 1 and 31 2 are formed by multilayer stacking of dielectric plates, first and second drivers composed of slot antennas can be formed by attaching driver elements 31 61 and 31 62 composed of a metal plate or metal film provided with a single-line slot or a cross-shaped slot to the inner surfaces of the bottom walls of each of sealed cases 31 1 and 31 2 and attaching ground conductors to the outer surfaces of each bottom wall. In addition, the first driver may be formed by connecting a probe to internal conductor 31 41 of coaxial con nection plug 31 31 shown in Fig. 19; and the second driver may be formed by connecting probes to internal conductors 31 42 and 31 43 of coaxial connection plug 31 32 and 31 33 shown in Fig. 20, the longitudinal direction of both probes being parallel to metal plate or metal film 31 62 and the portions extending in the longitudinal direction of both probes intersecting at the center point of metal plate or metal film 31 62 , this intersection being a right angle. In either of the above-described constructions of the first and second drivers installed in sealed cases 31 1 and 31 2 , respectively, the amount of phase shift between input and output is determined according to the relative angle of rotation between sealed cases 31 1 and 31 2 , and to rotate sealed cases 31 1 and 31 2 relative to each other, a construction may be adopted wherein sealed case 31 2 , coaxial cables 33 1 and 33 2 , and 90° 3-dB hybrid circuit 32 are fixed and sealed case 31 1 is caused to rotate; wherein sealed case 31 1 is fixed and sealed case 31 2 , coaxial cables 33 1 and 33 2 , and 90° 3-dB hybrid circuit 32 are caused to rotate as a unit; or wherein sealed case 31 1 and 90° 3-dB hybrid circuit 32 are fixed, coaxial cables 33 1 and 33 2 is formed from a flexible cable, and sealed case 31 2 is caused to rotate. The foregoing explanation relates to a case in which sealed cases 31 1 and 31 2 are both formed as cylinders with one closed end, but either sealed case 31 1 or 31 2 may be formed as a cylinder with one closed end in which the first or second driver is installed, and the other sealed case, i.e., sealed case 31 2 or 31 1 may be formed as a disk-shaped cover with the second or first driver attached to its inner surface and formed such that the cover rotatably fits with the end portion of the opening of the cylinder with one closed end. The foregoing explanation relates to a case in which 90° 3-dB hybrid circuit 32 is formed as a quarter-wave coupled-line directional coupler, but the hybrid circuit may also be formed as a branch line directional coupler. In addition, for cases in which the transmission fre quency bandwidth is relatively narrow, a two-branch terminal circuit may take the place of 90° 3-dB hybrid circuit 32, and the length of one of coaxial cables 33 1 and 33 2 , for example coaxial cable 33 2 , may be formed exactly one transmission quarter-wavelength longer than coaxial cable 33 1 . In this embodiment, the lengths of coaxial cable 33 1 and 33 2 must differ by exactly one transmission quarter-wavelength, but fabrication is relatively easy because the difference in length between coaxial cables 33 1 and 33 2 is fixed at one transmission quarter-wavelength. If the difference in lengths between coaxial cables 33 1 and 33 2 is selected to be one transmission quarter-wavelength for the center frequency of the transmission band, the difference in length between the coaxial cables will not precisely match the quarter-wavelength for frequencies outside the center frequency, but since this embodiment is intended for applications in which the transmission frequency band is relatively narrow, any operational error arising due to variance from the quarter-wavelength is minute and presents no practical problem. The foregoing explanation relates to a case in which the phase shifting device is constructed by assembling three-dimensional constituent elements, but the entire structure can be miniaturized by forming 90° 3-dB hybrid circuit 32 on a printed circuit board using a printed wiring method and forming coaxial cables 33 1 and 33 2 as microstrip wiring. The entire structure may also be made extremely compact and concise by providing a dielectric layer on the outer surface of sealed case 31 2 of coupler 31, and then employing a printed wiring method to form 90° 3-dB hybrid circuit 32 and microstrip wiring that takes the place of coaxial cables 33 1 and 33 2 . In a case in which a two-branch terminal circuit takes the place of 90° 3-dB hybrid circuit 32, the entire structure can be miniaturized by forming this component on a printed circuit board or on a dielectric layer provided on the outer surface of sealed cases 31 2 .

Claims: A phase shifting device comprising: two-partition circuit means (22) for partitioning input power into two substantially equal portions having a mutual phase difference of substantially 90°; a first microwave antenna means (21, 21 61 ) driven by two-partitioned output of said two-partition circuit means for generating circularly polarized waves and a second microwave antenna means (21, 21 62 ) for coupling circularly polarized waves generated by said first microwave antenna means, said first and second antenna means are installed as to be rotatable relative to each other around the axis joining the centers of both antennae; and a combining circuit means (24,27) for combining two outputs generated from circularly polarized waves coupled in said second microwave antenna means; characterized in, that an electrically conductive enclosure (21 1 , 21 2 ) is provided within which said first and second microwave antenna means (21 61 ,21 62 ) are installed, and having an inside diameter substantially equal to 0,285 λ 0 , where λ 0 is the free space wavelength corresponding to the designed frequency f 0 . A phase shifting device according to claim 1, wherein the electrically conductive enclosure (21 1 , 21 2 ) comprises first and second cylinders, each having one closed end for installing said first and second microwave antenna means (21 61 , 21 62 ) therein, the end portion of the opening of said first cylinder fitting with the end portion of the opening of said second cylinder, said first and second cylinders being rotatable to each other about their cylindrical axis. A phase shifting device according to claim 1, wherein the electrically conductive enclosure (21 1 or 21 2 ) comprises a cylinder with one closed end for installing therein either one of said first and second microwave antenna means (21 61 , 21 62 ) and a disk-shaped cover which fits with the end portion of the opening of said cylinder and upon the inner surface of which is attached the other of said first and second microwave antenna means, said cylinder and said cover being rotatable relative to each other about the cylindrical axis of said cylinder. A phase device according to one of claims 1 to 3, wherein said first and second microwave antenna means each comprise microstrip antennae (21 61 ; 21 62 ). A phase shifting device according to one of claims 1 to 3, wherein said first and second microwave antenna means each comprises slot antennae (21 71 ). A phase shifting device according to one of claims 1 to 3, wherein said first and second microwave antenna means (21 61 , 21 62 ) each comprise two probes provided in the same plane in an orthogonal relationship. A phase shifting device comprising: a first microwave antenna means (31 61 ) driven by input applied by way of an input terminal (31 31 ) for generating linearly polarized waves and a second microwave antenna means (31 62 ) for coupling two orthogonal components of linearly polarized waves generated by said first microwave antenna means, said first and second antenna means are installed as to be rotatable relative to each other around the axis joining the centers of both antennae; and a combining circuit means (32,33) for combining two outputs generated from two orthogonal components of linearly polarized waves coupled in said second microwave antenna means, characterized in, that an electrically conductive enclosure (31 1 , 31 2 ) is provided within which said first and second microwave antenna means (31 61 ,31 62 ) are installed, and having an inside diameter substantially equal to 0,285 λ 0 , where λ 0 is the free space wavelength corresponding to the designed frequency f 0 . A phase shifting device according to claim 7, wherein the electrically conductive enclosure (31 1 , 31 2 ) comprises first and second cylinders, each having one closed end for installing said first and second microwave antenna means (31 61 ,31 62 ) therein, the end portion of the opening of said first cylinder fitting with the end portion of the opening of said second cylinder, said first and second cylinders being rotatable to each other about their cylindrical axis. A phase shifting device according to claim 7, wherein the electrically conductive enclosure (31 1 or 31 2 ) comprises a cylinder with one closed end for installing therein either one of said first and second microwave antenna means (31 61 , 31 62 ) and a disk-shaped cover which fits with the end portion of the opening of said cylinder and upon the inner surface of which is attached the other of said first and second microwave antenna means, said cylinder and said cover being rotatable relative to each other about the cylindrical axis of said cylinder. A phase shifting device according to one of claims 7 to 9, wherein said first and second microwave antenna means each comprise microstrip antennas (31 61 ; 31 62 ). A phase shifting device according to one of claims 7 to 9, wherein said first and second microwave antenna means (31 61 ,31 62 ) each comprise slot antennas. A phase shifting device according to one of claims 7 to 9, wherein first microwave antenna means (31 61 ) comprises one probe, and said second microwave antenna means (31 62 ) comprise two probes provided in the same plane, extending portions of which are in an orthogonal relationship. A phase shifting device according to one of claims 7 to 9, wherein said combining circuit means (32,33) comprises a 90° 3-dB hybrid circuit (32). A phase shifting device according to one of claims 7 to 9, wherein said combining circuit means (32) comprises a two-branch terminal circuit.

Inventors: EBINE YOSHIO, KARIKOMI MASAHIRO, MATSUOKA TOHRU, EBINE, YOSHIO, KARIKOMI, MASAHIRO, MATSUOKA, TOHRU
Applicants: NIPPON DENGYO KOSAKU KK, NIPPON TELEGRAPH & TELEPHONE, NIHON DENGYO KOSAKU CO. LTD., NTT MOBILE COMMUNICATIONS NETWORK INC.
Application year: 1995
Main classification: H01P1
IPC codes: H01P1, H01Q3