SWITCHED-CAPACITOR ONE-BIT DIGITAL-TO-ANALOG CONVERTER WITH LOW SENSITIVITY TO OP-AMP OFFSET VOLTAGE

Abstract: A switched-capacitor DAC system includes an integrator circuit including an op amp (60) having an input lead (62, 64), an output lead (66, 68) and an integrator capacitor (C3, C4) connected between the input lead and the output lead. A sampling switch (78) is operable to connect an input capacitor (C1, C2) to be charged by an input voltage (V ref) during at least one of first and second nonoverlapping time intervals, wherein the first time interval is subdivided into first and second nonoverlapping sub-intervals and the second time interval is subdivided into third and fourth nonoverlapping sub-intervals. A transferring switch (80) is operable to connect the input capacitor to transfer charge from the input capacitor to transfer charge from the input capacitor to the integrator capacitor (C3, C4) during at least one of the first and third sub-intervals. A discharging switch (S8, S10) is operable to connect the input capacitor to a discharge node during at least one of the second and fourth sub-intervals. In a preferred embodiment of the present invention, the sampling switch connects the input capacitor during one of the first and second sub-intervals, the transferring switch connects the input capacitor during one of the first and third sub-intervals, and the discharging switch connects the input capacitor during one of the second and fourth sub-intervals.

Description: SWITCHED-CAPACI OR ONE-BIT DIGITAL-TO-ANALOG CONVERTER WITH LOW SENSITIVITY TO OP-AMP OFFSET VOLTAGE FIELD OF THE INVENTION The present invention relates to digital-to-analog converters ("DACs") and, more particularly, to switched-capacitor DACs including circuitry for reducing operational amplifier ("op amp") offset voltage nonlinearity errors. BACKGROUND OF THE INVENTION Sigma-delta analog-to-digital converters ("ADCε") and DACs have recently come into widespread use with the development of suitable process technology and the increase in digital audio and other applications. Sigma-delta converters utilize oversampling techniques (i.e., sampling at rates greater than the Nyquist rate) to achieve high signal-to-noise ratios. Such converters also exhibit excellent linearity. Additionally, sigma-delta converters are relatively straight-forward and inexpensive to implement due to their simplicity. A sigma-delta ADC converts an analog input voltage to digital output samples at a predetermined rate. A typical sigma-delta ADC includes a front-end modulator which, by performing an oversampling technique referred to as "noise shaping", manipulates the noise spectrum signal such that a major component of the quantization noise power is shifted to a frequency range higher than the upper frequency limit of the band of interest, which is typically the signal bandwidth (within the output samples). Subsequent filtering ("decimation") is performed in the digital domain to reduce the high frequency quantization noise component of the digital output samples. A conventional, first order sigma-delta ADC is shown in block diagram form in Fig. 1. The ADC includes a modulator 50 and a decimator 59. Modulator 50 converts an input voltage Vin, received on line 70, into a one-bit data stream X at a rate determined by the sampling frequency Kf . Modulator 50 performs oversampling and noise shaping on the input voltage. The one-bit data stream X is provided on line 57 to decimator 59 which low-pass filters the data stream to reduce the quantization noise component thereof, and provides filtered N-bit output samples at a rate fs on output line(s) 61. In short, the decimator 59 decimates, by a factor K, the one-bit data stream X. The modulator 50 includes an input circuit 51 which samples the input signal Vin and provides the sampled input signal to a summing circuit 54. Summing circuit 54 subtracts a feedback signal (described below), received on line 65, from the sampled input signal and provides the output signal difference on line 55 to an integrator 56. Integrator 56 conventionally "integrates" the output signal difference from the summing circuit 55 and provides an output signal A to a clocked, latched comparator 58. Summing circuit 54 can generally be considered as an input section of the integrator. Comparator 58 conventionally is clocked at a rate Kf S by a clock signal applied to line 58A. For each clock pulse, comparator 58 "compares" signal A to ground and provides a one-bit output. The sequence of outputs for a sequence of clock pulses form on line 57 the one-bit data stream output X of the modulator. Thus, comparator 58 is effectively a one-bit ADC. Data stream X is also provided on feedback line 63 to a feedback circuit 52, which includes a one-bit DAC (not shown) . The stream X controls a switch 53 within the feedback circuit 52 such that either a positive feedback reference voltage +Vref, received on line 74, or a negative feedback reference voltage -Vref, received on line 76, will be applied, for each bit in stream X, via feedback line 65, to circuitry 66, which generates and supplies a corresponding feedback signal to the summing circuit 54. This switching operation of the feedback circuit 52 of the modulator 50 is conventional for a closed-loop circuit and should be readily understood by those skilled in the art. Sigma-delta modulators are typically implemented with εwitched-capacitor circuits. While the operation of the modulator was generally described above in terms of delivering signals, in a switched-capacitor implementation, the signals are packets of charge. Thus, a charge packet is developed each time the input signal is sampled and each time a feedback signal is generated, with such charges or the net charge being "transferred" from the input and feedback circuits to the integrator via the summing circuit. In a switched-capacitor implementation, the integrator 50 typically includes an op amp and a feedback-connected integrator capacitor (not shown in Fig. 1). The input and feedback circuits typically include switches and capacitors for respectively sampling the input and feedback reference voltages. During operation, an input capacitor is typically charged by a voltage source (i.e., the input voltage source or feedback reference voltage source) through a first switch (or switches) during a first time interval (clock phase), and charge is thereafter "transferred" through a second switch (or switches) from the input capacitor to the feedback-connected integrator capacitor during a second, nonoverlapping time interval. The switches often include CMOS transistors due to their high performance and yield. It should be appreciated by those skilled in the art that, as used in the art and herein, charge "transfer" refers to a charging of the integrator capacitor C3 by the integrator output voltage to compensate for charging (sampling) of the input capacitor Cl due to the equipotential surface between one plate of the input capacitor Cl and one plate of the integrator capacitor C3 (see Fig. 2). Thus, a literal, physical movement of charge from the input capacitor to the integrator capacitor may not occur. The above description of the modulator, in which a single input voltage is sampled (with respect to ground) , assumes the integrator 11 based on use of a single-ended op amp. As will be understood by those skilled in the art, however, the integrator may include a differential amplifier for which positive and negative input voltages are separately sampled. Whether single-ended or differential, CMOS op amps typically have associated input offset voltages within the range of 1-10 mv (whereas ideally the input offset voltage should be 0). During operation, the difference between the voltages on the input terminals of a differential op amp will be equal to the input offset voltage, when the output voltage is at 0 volts. In sigma-delta ADC modulators, the input offset voltage of the op amp may charge the feedback circuit capacitor(s) (i.e., the offset voltage is sampled by the feedback circuit) and that charge may be "transferred" to the integrator. Ideally, the only charge transferred from the feedback circuit should be that due to the sampled feedback reference voltages. As will be appreciated by those skilled in the art, the amount of charge transferred from the feedback circuit to the integrator is typically controlled by the transition density (i.e., the number of transitions from 1 to 0 or from 0 to 1) of the digital output signal (X in Fig. 1) of the modulator. Because the transition density of the digital output signal is typically nonlinear, the transfer of charge resulting from sampling the input offset voltage is also nonlinear. As a result of the nonlinear transfer of charge due to the sampling of the input offset voltage, such modulators may suffer from integral nonlinearity performance inaccuracies and encounter repetitive noise patterns known as "idle tones". The following is a detailed discussion of such prior art modulator performance. Referring to Fig. 2, where like elements are referred to by identical reference characters to those of Fig. 1, a prior art switched-capacitor DAC system is shown including a DAC 52 and an integrator 56. During operation, the DAC 52 samples the reference voltage Vref and transfers charge corresponding to +Vref or -Vref to the integrator 56. The reference voltage Vref can be considered an input voltage to the DAC. Integrator 56 includes an op amp 60, a first integrator capacitor C3 connected between noninverting output lead 66 and inverting input lead 62 of the op amp, and a second integrator capacitor C4 connected between inverting output lead 68 and noninverting input lead 64. The switched-capacitor DAC 52 includes input lines 74 and 76 respectively receiving the positive and the negative terminals of the reference voltage Vref. A first input capacitor Cl is coupled to the input lines 74 and 76 through a first switching circuit 78 and to the input leads of the op amp through a second switching circuit 80. A second input capacitor C2 is similarly coupled to the input lines 74 and 76 through the first switching circuit 78 and to the input leads 62 and 64 of the op amp through the second switching circuit 80. Capacitors Cl and C2 sample (i.e., are charged by) the reference voltage Vref through switching circuit 78 and transfer charge to capacitors C3 and C4 through switching circuit 80. The values of capacitors Cl and C2 are preferably equal as are the values of capacitors C3 and C4. Switching circuit 78 includes switches labeled with the symbol PI or P2. The switches labeled PI are controlled by the control signal PI and the switches labeled P2 are controlled by the control signal P2 (see Fig. 3). Switching circuit 80 includes switches labeled with the symbol Rl or R2. The switches labeled Rl are controlled by the control signal Rl and the switches labeled R2 are controlled by the control signal R2 (see Fig. 3). Shown in the timing diagram of Fig. 3 are the control signals PI, P2, Rl and R2 as well as the digital input signal Y used to generate signals Rl and R2. In a sigma-delta ADC modulator application, signal Y is typically the digital output signal (i.e., signal X in Fig. 1). The control signals are shown on the same time axis and the vertical placement of one above the other does not signify that one attains different voltage levels than the other; the "high" and "low" voltage levels of the signals are relative to each other only. As is conventional for a switched-capacitor circuit, the PI and P2 switches operate in two nonoverlapping time intervals (or clock phases). During interval 1 (marked on the time axis), signal PI is at a high voltage level and signal P2 is at a low voltage level. During interval 2, signal PI is low and signal P2 is high. Signal PI controls the switches labeled PI such that during interval 1 (when PI is high) , the PI labeled switches are closed (and conduct current) and, during interval 2 (when PI is low), the PI labeled switches are open (and prevent the flow of current). Conversely, the switches labeled P2 are open during interval 1 and are closed during interval 2. It is important that the signals PI and P2 are not high at the same time so that accurate reference voltage sampling occurs. Thus, as will be understood by those skilled in the art, the circuit (not shown) generating signals PI and P2 typically establishes a "break-before-make" operation to ensure that the control signals are not simultaneously high. The control signals Rl and R2 depend on the level of the signal Y in accordance with the following relationships. Rl = P1»YB + P2»Y and R2 = Pl»Y + P2»YB, where YB is the complement of Y, "+" represents a logical OR operation and "•" represents a logical AND operation. When Y is low, signal Rl is identical to signal PI and signal R2 is identical to signal P2. When Y is high, signal Rl is identical to signal P2 and signal R2 is identical to signal PI. Referring back to Fig. 2, switching circuit 78 includes a first switch SI connected between input line 74 and the left plate of capacitor Cl, a second switch S2 connected between input line 76 and the left plate of capacitor Cl, a third switch S3 connected between input line 76 and the left plate of capacitor C2, and a fourth switch S4 connected between input line 74 and the left plate of capacitor C2. Switching circuit 80 includes a first switch S5 connected between the right plate of capacitor Cl and the inverting input lead 62 of op amp 60, a second switch S6 connected between the right plate of capacitor Cl and the noninverting input lead 64, a third switch S7 connected between the right plate of capacitor C2 and the noninverting input lead 64, and a fourth switch S8 connected between the right plate of capacitor of C2 and the inverting input lead 62. As labeled, switches SI and S3 are controlled by control signal PI, switches S2 and S4 are controlled by control signal P2, switches S6 and S8 are controlled by control signal Rl, and switches S5 and S7 are controlled by control signal R2. As should be readily understood by those skilled in the art, the input capacitors Cl and C2 operate to sample the reference voltages through switching circuit 78 and to transfer charge to the integrating capacitors C3 and C4 through the switching circuit 80. The cross-coupled arrangement of the input capacitors enables the reference voltages to be sampled during both time intervals and charge to be transferred during both time intervals. The timing diagram of Fig. 3 assumes that signal Y transitions once during intervals 1 and 2 and that the transition occurs between interval 2 and interval 1. While the prior DAC of Fig. 2 operates effectively to sample the reference voltages during both time intervals and to transfer charge to the integrating capacitors during both time intervals, the DAC may produce integral nonlinearity errors due to the presence of an op amp input offset voltage. Fig. 4 schematically illustrates the prior art DAC of Fig. 2 for the situation where the reference voltages are equal to 0 volts (i.e., the input lines 74 and 76 are tied to ground) and the op amp 70 has an associated input offset voltage V QS , so that the effect of the input offset voltage can be more easily analyzed. Such an analysis follows. During interval 1, assuming signal Y is low, the left plate of capacitor Cl is connected to ground through switch SI and the right plate of capacitor Cl is connected through switch S6 to the noninverting input lead 64 which is at a voltage level of -V nς /2. [The common-mode voltage of the input leads of the op amp is assumed to be 0 volts. Therefore, the inverting input lead 62 is at a voltage level of +V /2 and the noninverting input lead 64 is at a voltage level of -V Qς /2. ] During the immediately following interval 2, assuming that Y does not transition from low to high, the left plate of capacitor Cl is connected to ground through switch S2 and the right plate of capacitor Cl is connected through switch S5 to inverting input lead 62 (which is at a voltage level of +V /2) . Thus, a total charge (related to the change in voltage on the right plate of capacitor Cl) of Cl»V Qq has accumulated on capacitor Cl during the two intervals and that same charge of C1«V 0 _ is transferred to capacitor C3. As will be appreciated by those skilled in the art, an equal and opposite charge will be transferred from capacitor C2 to capacitor C4 due to the differential arrangement of the circuitry and the capacitors Cl and C2 being of equal value. Therefore, the charging of capacitor C2 and the transfer of charge from capacitor C2 to C4 will not be separately analyzed. In sum, if signal Y remains low (does not transition), a charge equal to C1«V Q is transferred to the integrator capacitor C3 during the second of two consecutive intervals. However, when signal Y transitions between two consecutive intervals, the right hand plate of the input capacitor remains connected to the same node and no charge is transferred, as described below. During interval 2 (assuming Y is low) , the left plate of capacitor Cl is connected through switch S2 to ground and the right plate of capacitor Cl is connected through switch S5 to the inverting input lead 62 (at a voltage level of +V /2). During the subsequent interval 1 (assuming Y transitions to high), the left plate of capacitor Cl is connected to ground through switch SI and the right plate of capacitor Cl remains connected through switch S5 to the inverting input lead 62. Thus, the voltage on the right plate of capacitor Cl does not change between intervals 2 and 1 and no charge is transferred. In sum, if Y does not transition between two consecutive time intervals, then a charge of Cl»V Qq is transferred and if Y does transition between the time intervals, then no charge is transferred. The total charge transferred due to the input offset voltage of the op amp over a period of time, therefore, depends on the density of transitions in the signal Y. In a sigma-delta ADC modulator, the transition density in the output signal (Y) is typically nonlinear. Fig. 5 is a graph illustrating the transition density for a digital output signal of a second-order sigma-delta ADC modulator over an analog input voltage range of -1 volt to +1 volt. As shown, the transition density is approximately equal to 70 percent for mid-scale code (i.e., 0 volts) and is approximately equal to 25 percent for positive and negative full-scale code (i.e., +1 volts). As a result, a sigma-delta ADC modulator employing the prior art DAC of Fig. 2 may encounter integral nonlinearity accuracy problems and idle tones. Accordingly, a general object of the present invention is to provide a high-performance switched-capacitor DAC including circuitry for reducing op amp offset voltage nonlinearity errors. Other objects and advantages will be apparent from the detailed description below. SUMMARY OF THE INVENTION The aforementioned drawbacks of the prior art are overcome by a method and apparatus of the present invention in which a switched-capacitor input circuit charges an input capacitor with an input voltage during sampling time intervals and transfers charge from the input capacitor to an integrator capacitor during transferring time sub-intervals. One plate of the input capacitor is connected through a transferring switch to an input lead of the op amp during the transferring time sub-intervals. That plate of the input capacitor is connected through a discharge switch to a discharge node during discharging time sub-intervals to discharge offset voltage charge accumulated on the input capacitor during a previous transferring sub-interval. More particularly, in accordance with the preferred embodiment of the present invention, a switched-capacitor DAC system includes an integrator circuit including an op amp having an input lead, an output lead and an integrator capacitor connected between the input lead and the output lead. A switched-capacitor DAC includes an input capacitor connectable through a sampling switch to be charged by an input voltage at a sample rate and connectable through a transferring switch to transfer charge from the input capacitor to the integrator capacitor at a transfer rate. The DAC further includes a discharge switch for connecting the input capacitor to a discharge node at a predetermined rate related to the transfer rate such that any charge transfer due to op amp offset voltage will be substantially independent of any nonlinearities in the transfer rate. In one embodiment of the present invention, the input capacitor is charged by the input voltage during first and second nonoverlapping time intervals. The first time interval is subdivided into first and second nonoverlapping sub-intervals and the second time interval is subdivided into third and fourth nonoverlapping sub-intervals. Charge is transferred from the input capacitor to the integrator capacitor during the first and third sub-intervals, and the input capacitor is connected to the discharge node during either the second or fourth sub-intervals. In another embodiment of the present invention, the input capacitor is connected to the discharge node during both the second and fourth sub-intervals. BRIEF DESCRIPTION OF THE DRAWINGS Fig. 1 is a block diagram of a prior art sigma-delta ADC system; Fig. 2 is a schematic diagram of a prior art switched- capacitor DAC system; Fig. 3 is a timing diagram of the control signals which control operation of the switches within the DAC system of Fig. 2; Fig. 4 is a schematic diagram of the DAC system of Fig. 2 illustrating the op amp input offset voltage for a particular DAC input voltage; Fig. 5 is a graph of the transition density of a typical digital output signal of a second-order sigma-delta ADC modulator; Fig. 6 is a schematic diagram of a DAC system according to the present invention; Fig. 7 is a timing diagram of the control signals which control operation of the switches within the DAC system of Fig. 6; Fig. 8 is a schematic diagram of the DAC system of Fig. 6 illustrating the op amp input offset voltage for a particular DAC input voltage; Fig. 9 is an alternate embodiment timing diagram of the control signals which control operation of the switches within the DAC system of Fig. 6; and Fig. 10 is a schematic diagram of an alternate embodiment DAC system according to this invention. DETAILED DESCRIPTION By contrast with prior art switched-capacitor DAC systems in which the amount of input offset voltage charge transferred from an input capacitor to an integrating capacitor is dependent upon the transition density of the input control signal which is used to generate the control signals for controlling the transfer switches, the amount of charge (due to the input offset voltage) transferred from the input capacitor to the integrating capacitor in the DAC system of the present invention is independent of the transition density of the input control signal. In a preferred embodiment of the present invention, one plate of the input capacitor is periodically connected to a discharge node at a known voltage (e.g., ground), to discharge charge build-up on the input capacitor due to op amp input offset voltage before charge is transferred from the input capacitor to the integrating capacitor. Fig. 6 is a schematic diagram of a first embodiment of a switched-capacitor DAC circuit according to the present invention which ensures that the amount of charge transferred due to the input offset voltage will be independent of the transition density of digital input signal Y. Thus, the DAC circuit of the present invention is immune to the nonlinear transition density of signal Y. The DAC circuit of Fig. 6 is identical to the prior art DAC circuit of Fig. 2 except for the addition of switches S9 and S10. Switch S9 is connected between the right plate of capacitor Cl and a reference voltage such as ground, and switch S10 is connected between the right plate of plate of capacitor C2 and the same reference voltage. The operation of switches S9 and S10 is controlled by the periodic control signal P3, shown in the timing diagram of Fig. 7. Switch S9 operates to periodically connect the right plate of capacitor Cl to ground to discharge charge accumulation on capacitor Cl due to the input offset voltage of the op amp. Switch S10 operates similarly with respect to capacitor C2. As shown in the timing diagram of Fig. 7, each of the intervals 1 and 2 is subdivided into two sub-intervals. Interval 1 includes sub-intervals A and B, and interval 2 includes sub-intervals C and D. Control signals Rl and R2 are governed by the following equations. Rl = P1S«YB + P2S-Y and R2 = P1S»Y + P2S-YB, where PIS is similar to signal PI but with a shortened high duration and signal P2S is similar to signal P2 but with the εame shortened high duration. If signal Y is low, signal Rl is high for sub-interval A and is low for sub-interval B and signal P3 is low during sub-interval A and is high during sub-interval B. Similarly, if Y is low, signal R2 is high during sub-interval C and is low during sub-interval D and signal P3 is low during sub-interval C and is high during sub-interval D. If Y is high, signals Rl and R2 are reversed. It is important that signal P3 is not simultaneously high with signal Rl or R2. Thus, the circuit (not shown) which generates such control signals applies appropriate break-before-make operations. Fig. 8 illustrates the switched capacitor DAC of the present invention having input voltages of 0 volts (i.e., with the input lines tied to ground) and an operational amplifier input offset voltage V Qς , so that the effect of the input offset voltage can be more easily analyzed. Such an analysis follows. During interval 2, sub-interval C, assuming signal Y is low, the left plate of capacitor Cl is connected to ground through switch S2 and the right plate of capacitor Cl is connected through switch S5 to the inverting input lead 62. The right plate of capacitor Cl is therefore at a voltage level of + os /2• Thus, capacitor Cl will charge to a value of -C1»V /2. During interval 2, sub-interval D, the left plate of capacitor Cl is still connected through switch S2 to ground and the right plate of capacitor Cl is connected through switch S9 to ground. Thus, the charge of -C1«V /2 on capacitor Cl will be discharged. During the subsequent interval 1, sub-interval A, assuming signal Y does not transition, the left plate of capacitor Cl will be connected through switch SI to ground and the right plate of capacitor Cl will be connected through switch S6 to the noninverting input lead 64. The right plate of capacitor Cl will therefore be at a voltage level of -V QS /2 and capacitor Cl will charge to Cl«V_ ς /2. Thus, a charge of Cl»V 0 _/2 will be transferred to integrating capacitor C4 (and due to the differential arrangement of the op amp and DAC circuit, an equal and opposite charge of -C1«V /2 will be transferred from input capacitor C2 to integrating capacitor C3) . If, instead, signal Y had transitioned between interval 2 and interval 1, the same charge would be transferred during interval 1, as discussed below. The charge on capacitor Cl would still be discharged during interval 2, sub-interval D. However, during the subsequent interval 1, sub-interval A, the right plate of capacitor Cl would be connected through switch S5 to the inverting input lead 62. The right plate of capacitor Cl would therefore be at a voltage level of +V Qg /2 and capacitor Cl would charge to -Cl»V oq /2. Thus, a charge of -Cl»V QS /2 would be transferred to capacitor C3 (with an equal and opposite charge of C1«V 0 /2 being transferred to capacitor C4) which is the same as that transferred without signal Y transitioning (discussed above). Thus, the only charge (due to the input offset voltage) transferred during a transferring time sub-interval is that sampled during the same transferring sub-interval. The circuit operates to discharge charge accumulation (due to the input offset voltage) from a previous transferring time sub-interval so that the input capacitors have no "memory" of the previous transferring sub-interval . In sum, as shown and described, the amount of charge (due to the input offset voltage) transferred from the input capacitors to the integrating capacitors is independent of the transition density of the control signal Y. Accordingly, a sigma-delta ADC modulator employing the DAC circuit of the present invention would not suffer from the offset voltage nonlinearity problems of the prior art. Assuming that the control signal Y can transition only once during intervals 1 and 2 (which is typical for a sigma-delta ADC modulator), it is only necessary to close switches S9 and S10 once before each possible transition of signal Y. Thus, an alternate embodiment timing diagram of the control signals is shown in Fig. 9. As shown, signal P3 is only high once during intervals 1 and 2 (i.e., in sub-interval D) before the transition of signal Y. As will be appreciated by those skilled in the art, however, the phase of the control signal Y may vary with respect to the control signals PI and P2, depending on the application and the switched-capacitor circuit. Thus, the timing diagram of the control signal P3 can be varied accordingly, provided that the control signal P3 goes high (i.e., switches controlled by P3 are closed) once before each possible transition of signal Y. Additionally, as those skilled in the art will appreciate, for a "chopper-stabilizer" DAC circuit, in which the connections of the input leads of the op amp are interchanged (reversed) during each interval in order to cancel completely the effect of the input offset voltage, the timing diagram of Fig. 7 must be used to effectuate that cancelling. Fig. 10 is a schematic diagram of an alternate embodiment of the DAC circuit of the present invention. In the embodiment of Fig. 10, the right plate of capacitor Cl is connected through switch 11 to the right plate of capacitor C2 and switch Sll is controlled by control signal P3. Assuming that the value of capacitor Cl is equal to the value of capacitor C2, the charge accumulated on capacitor Cl due to the input offset voltage will be equal and opposite to that accumulated on capacitor C2. Therefore, when the right plates of the capacitors are connected together through switch 11, the charge on the capacitors will effectively cancel one another. Thus, the circuit of Fig. 10 effectively operates identically to that of Fig. 6. As those skilled in the art will readily appreciate, however, the advantage of the Fig. 6 embodiment, in which the right plate of the capacitors are connected to a fixed voltage, is that the fixed voltage serves as a common-mode reference voltage for the op amp and will aid in dynamically correcting any tendency for the common-mode voltage to drift. While there have been shown and described what are at present considered the preferred embodiments for the present invention, which have been disclosed by way of example only, it would be obvious to those skilled in the art that various changes and modifications can be made therein without departing from the spirit and scope of the invention as presented above and as defined by the appended claims and equivalents thereto. Particularly, while specific switching arrangements and timing control patterns have been disclosed, others are envisioned which effectuate a transfer of offset voltage charge independent of any nonlinearities in the input control signal. In addition, while the DAC of the present invention has been shown an described as being used in a sigma-delta ADC application which includes an integrator, the DAC of of the present invention may alternatively be connected to a switched-capacitor gain stage.

Claims: CL IMS 1. A switched-capacitor DAC system for use with a sigma-delta ADC modulator having an integrator circuit including an op amp having an input lead, an output lead and an integrator capacitor connected between the input lead and the output lead, the DAC comprising: an input capacitor; a sampling switch operable to connect the input capacitor to be charged by an input voltage during at least one of first and second nonoverlapping time intervals, wherein the first time interval is subdivided into first and second nonoverlapping sub-intervals and the second time interval is subdivided into third and fourth nonoverlapping sub-intervals; a transferring switch operable to connect the input capacitor to transfer charge from the input capacitor to the integrator capacitor during at least one of the first and third sub-intervals; and a discharging switch operable to connect the input capacitor to a discharge node during at least one of the second and fourth sub-intervals. 2. A DAC as claimed in claim 1, wherein the discharging switch connects the input capacitor during one of the second and fourth sub-intervals. 3. A DAC as claimed in claim 2, wherein the discharging switch connects the input capacitor during both the second and fourth sub-intervals. 4. A switched-capacitor DAC system for use with a sigma-delta ADC modulator having an integrator circuit including an op amp having an input lead, an output lead and an integrator capacitor connected between the input lead and the output lead, the DAC comprising: an input capacitor; sampling switch means for connecting the input capacitor to be charged by an input voltage during at least one of first and second nonoverlapping time intervals, wherein the first time interval is subdivided into first and second nonoverlapping sub-intervals and the second time interval is subdivided into third and fourth nonoverlapping sub-intervals; transferring switch means for connecting the input capacitor to transfer charge from the input capacitor to the integrator capacitor during at least one of the first and third sub-intervals; and discharging switch means for connecting the input capacitor to a discharge node during at least one of the second and fourth sub-intervals. 5. A DAC as claimed in claim 4, wherein the discharging switch means connects the input capacitor during one of the second and fourth sub-intervals. 6. A DAC as claimed in claim 4, wherein the discharging switch means connects the input capacitor during both the second and fourth sub-intervals. 7. A method for converting a digital signal to an analog signal comprising the steps of: connecting the input capacitor to be charged by an input voltage during at least one of first and second nonoverlapping time intervals, wherein the first time interval is subdivided into first and second nonoverlapping sub-intervals and the second time interval is subdivided into third and fourth nonoverlapping sub-intervals; connecting the input capacitor to transfer charge from the input capacitor to an integrator capacitor during at least one of the first and third sub-intervals; and connecting the input capacitor to a discharge node during at least one of the second and fourth sub-intervals. 8. A method as claimed in claim 7 wherein the step of connecting the input capacitor to a discharge node includes the step of connecting the input capacitor to a discharge node during one of the second and fourth sub-intervals. 9. A method as claimed in claim 8 wherein the step of connecting the input capacitor to a discharge node includes the step of connecting the input capacitor to a discharge node during both the second and fourth sub-intervals. 10. A switched-capacitor DAC system, the DAC system coupled to a switched-capacitor gain circuit having a gain capacitor, the DAC system comprising: an input capacitor; a sampling switch operable to connect the input capacitor to be charged by an input voltage during at least one of first and second nonoverlapping time intervals, wherein the first time interval is subdivided into first and second nonoverl pping sub-intervals and the second time interval is subdivided into third and fourth nonoverlapping sub-intervals; a transferring switch operable to connect the input capacitor to transfer charge from the input capacitor to the gain capacitor during at least one of the first and third sub-intervals; and a discharging switch operable to connect the input capacitor to a discharge node during at least one of the second and fourth sub-intervals.

Inventors: MCCARTNEY DAMIEN, MCCARTNEY, DAMIEN
Applicants: ANALOG DEVICES INC, ANALOG DEVICES, INC.
Application year: 1995
Main classification: H03M3
IPC codes: H03M3