OBJECT POSITION DETECTOR

Abstract: A proximity sensor system includes a sensor matrix array having a characteristic capacitance on horizontal and vertical conductors connected to sensor pads. The capacitance changes as a function of the proximity of an object or objects to the sensor matrix. The change in capacitance of each node in both the X and Y directions of the matrix due to the approach of an object is converted to a set of voltages in the X and Y directions. These voltages are processed by digital circuitry to develop electrical signals representative of the centroid of the profile of the object, i.e., its position in the X and Y dimensions. Noise reduction and background level setting techniques inherently available in the architecture are employed.

Description: SEECIEICΔIIΩE Title Of The Invention Object Position Detector Background Of The Invention 1. Field Of The Invention The present invention relates to object position sensing transducers and systems. More particularly, the present invention relates to object position recognition useful in applications such as cursor movement for computing devices and other applications. 2. The Prior Art Numerous devices are available or have been proposed for use as object position detectors for use in computer systems and other applications. The most familiar of such devices is the computer "mouse". While extremely popular as a position indicating device, a mouse has mechanical parts and requires a surface upon which to roll its position ball. Furthermore, a mouse usually needs to be moved over long distances for reasonable resolution. Finally, a mouse requires the user to lift a hand from the keyboard to make the cursor movement, thereby upsetting the prime purpose, which is usually typing on the computer. Trackball devices are similar to mouse devices. A major difference, however is that, unlike a mouse device, a trackball device does not require a surface across which it must be rolled. Trackball devices are still expensive, have moving parts, and require a relatively heavy touch as do the mouse devices. They are also large in size and doe not fit well in a volume- sensitive application like a laptop computer. There are several available touch-sense technologies which may be employed for use as a position indicator. Resistive-membrane position sensors are known and used in several applications. However, they generally suffer from poor resolution, the sensor surface is exposed to the user and is thus subject to wear. In addition, resistive-membrane touch sensors are relatively expensive. A one-surface approach requires a user to be grounded to the sensor for reliable operation. This cannot be guaranteed in portable computers. An example of a one-surface approach is the UnMouse product by MicroTouch, of Wilmington, MA A two-surface approach has poorer resolution and potentially will wear out very quickly in time. Resistive tablets are taught by United States Patent No. 4,680,430 to Yoshikawa, United States Patent No. 3,497,617 to Ellis and many others. The drawback of all such approaches is the high power consumption and the high cost of the resistive membrane employed. Surface Acoustic Wave (SAW) devices have potential use as position indicators. However, this sensor technology is expensive and is not sensitive to light touch. In addition, SAW devices are sensitive to residue buildup on the touch surfaces and generally have poor resolution. Strain gauge or pressure plate approaches are an interesting position sensing technology, but suffer from several drawbacks. This approach may employ piezo-electric transducers. One drawback is that the piezo phenomena is an AC phenomena and may be sensitive to the user's rate of movement. In addition, strain gauge or pressure plate approaches are somewhat expensive because special sensors are required. Optical approaches are also possible but are somewhat limited for several reasons. All would require light generation which will require external components and increase cost and power drain. For example, a "finger-breaking" infra-red matrix position detector consumes high power and suffers from relatively poor resolution. There have been numerous attempts to provide a device for sensing the position of a thumb or other finger for use as a pointing device to replace a mouse or trackball. Desirable attributes of such a device are low power, low profile, high resolution, low cost, fast response, and ability to operate reliably when the finger carries electrical noise, or when the touch surface is contaminated with dirt or moisture. Because of the drawbacks of resistive devices, many attempts have been made to provide pointing capability based on capacitively sensing the position of the finger. United States Patent No. 3,921,166 to Volpe teaches a capacitive matrix in which the finger changes the transcapacitance between row and column electrodes. United States Patent No. 4,103,252 to Bobick employs four oscillating signals to interpolate x and y positions between four capacitive electrodes. United States Patent No.4,455,452 to Schuyler teaches a capacitive tablet wherein the finger attenuates the capacitive coupling between electrodes. United States Patent No. 4,550,221 to Mabusth teaches a capacitive tablet wherein the effective capacitance to "virtual ground" is measured by an oscillating signal. Each row or column is polled sequentially, and a rudimentary form of interpolation is applied to resolve the position between two rows or columns. An attempt is made to address the problem of electrical interference by averaging over many cycles of the oscillating waveform. The problem of contamination is addressed by sensing when no finger was present, and applying a periodic calibration during such no-finger-present periods. United States Patent No. 4,639,720 to Rympalski teaches a tablet for sensing the position of a stylus. The stylus alters the transcapacitance coupling between row and column electrodes, which are scanned sequentially. United States Patent No. 4,736,191 to Matzke teaches a radial electrode arrangement under the space bar of a keyboard, to be activated by touching with a thumb. This patent teaches the use of total touch capacitance, as an indication of the touch pressure, to control the velocity of cursor motion. Pulsed sequential polling is employed to address the effects of electrical interference. United States Patent Nos. 4,686,332 and 5,149,919, to Greanias, teaches a stylus and finger detection system meant to be mounted on a CRT. As a finger detection system, it's X/Y sensor matrix is used to locate the two matrix wires carrying the maximum signal. With a coding scheme these two wires uniquely determine the location of die finger position to the resolution of the wire stepping. For stylus detection, Greanias first coarsely locates it, then develops a virtual dipole by driving all lines on one side of the object in one direction and all lines on die opposite side in the opposite direction. This is done three times with different dipole phases and signal polarities. Assuming a predetermined matrix response to the object, the three measurements present a set of simultaneous equations that can be solved for position. United States Patent No. 4,733,222 to Evans is the first to teach a capacitance touch measurement system that interpolates to a high degree. Evans teaches a three terminal measurement system that uses a drive, sense and electrode signal set (3 signals) in its matrix, and bases the measurement on die attenuation effect of a finger on d e electrode node signal (uses a capacitive divider phenomena). Evans sequentially scans through each drive set to measure the capacitance. From the three largest responses an interpolation routine is applied to determine finger position. Evans also teaches a zeroing technique tiiat allows "no-finger" levels to be cancelled out as part of the measurement. United States Patent No. 5,016,008 to Gruaz describes a touch sensitive pad that also uses interpolation. Gruaz uses a drive and sense signal set (2 signals) in d e touch matrix and like Evans relies on the attenuation effect of a finger to modulate the drive signal. The touch matrix is sequentially scanned to read the response of each matrix line. An interpolation program then selects the two largest adjacent signals in both dimensions to determine the finger location, and ratiometrically determines the effective position from those 4 numbers. Gerpheide, PCT application US90/04584, publication No. W091/03039, United States Patent No. 5,305,017 applies to a touch pad system a variation of die virtual dipole approach of Greanias. Gerpheide teaches die application of an oscillating potential of a given frequency and phase to all electrodes on one side of the virtual dipole,and an oscillating potential of the same frequency and opposite phase to those on the other side. Electronic circuits develop a "balance signal" which is zero when no finger is present, and which has one polarity if a finger is on one side of die center of the virtual dipole, and the opposite polarity if die finger is on die opposite side. To acquire die position of the finger initially, the virtual dipole is scanned sequentially across the tablet. Once the finger is located, it is "tracked" by moving die virtua dipole toward die finger once the finger has moved more than one row or column. Because the virtual dipole method operates by generating a balance signal that is zero when die capacitance does not vary with distance, it only senses the perimeter of d e finge contact area, rather than the entire contact area. Because d e method relies on synchronous detection of die exciting signal, it must average for long periods to reject electrical interference, an hence it is slow. The averaging time required by this method, together with die necessity to search sequentially for a new finger contact once a previous contact is lost, makes this method, like tiios before it, fall short of the requirements for a fast pointing device mat is not affected by electrical interference. It should also be noted that all previous touch pad inventions that used interpolation placed rigorous design requirements on their sensing pad. Greanias and Evans use a complicated and expensive drive, sense and electrode line scheme to develop their signal. Gruaz and Gerpheide use a two signal drive and sense set. In the present invention the driving and sensing is done on the same line. This allows the row and column sections to be symmetric and equivalent. This in turn allows independent calibration of all signal paths, which makes board layout simpler and less constraining, and allows for more unique sensor topologies. The shortcomings of the inventions and techniques described in die prior art can also be traced to the use of only one set of driving and sensing electronics, which was multiplexed sequentially over the electrodes in the tablet. This arrangement was cost effective in die days o discrete components, and avoided offset and scale differences among circuits. The sequential scanning approach of previous systems also made them more susceptible to noise. Noise levels could change between successive measurements, thus changing the measured signal and the assumptions used in interpolation routines. Finally, all previous approaches assumed a particular signal response for finger position versus matrix position. Because the transfer curve is very sensitive to many parameters and is not a smooth linear curve as Greanias and Gerpheide assume, such approaches are limited in die amount of interpolation they can perform. In prior co-pending application serial No. 08/115,743, filed August 31, 1993, now United States Patent No. , a two-dimensional capacitive sensing system equipped with a separate set of drive/sense electronics for each row and for each column of a capacitive tablet is disclosed. All row electrodes are sensed simultaneously, and all column electrodes are sensed simultaneously. The sensed signals are processed by analog circuitry. It is thus an object of the present invention to provide a two-dimensional capacitive sensing system equipped with a separate set of drive/sense electronics for each row and for each column of a capacitive tablet, wherein all row electrodes are sensed simultaneously, and all column electrodes are sensed simultaneously. It is a further object of the present invention to provide an electronic system tiiat is sensitive to the entire area of contact of a finger or other conductive object with a capacitive tablet, and to provide as output die coordinates of some measure of the center of this contact area while remaining insensitive to the characteristic profile of the object being detected. It is a further object of die present invention to provide an electronic system that provides as output some measure of area of contact of a finger or other conductive object with a capacitive tablet Yet another object of the present invention is to provide a two-dimensional capacitive sensing system equipped with a separate set of drive sense electronics for each row and for each column of a capacitive tablet, wherein all row electrodes are sensed simultaneously, and all column electrodes are sensed simultaneously and wherein die information defining the location of a finger or other conductive object is processed in digital form. It is a further object of the present invention to provide a two-dimensional capacitive sensing system wherein all row electrodes are sensed simultaneously, and all colum electrodes are sensed simultaneously and wherein digital electronic processing is employe efficiendy to define the location of a finger or other conductive object Brief Description Of The Invention With the advent of very high levels of integration, it has become possible t integrate many channels of driving/sensing electronics into one integrated circuit, along with th control logic for operating d em, and the interface electronics to allow the pointing device t communicate direcdy with a host microprocessor. The present invention uses adaptive analo techniques to overcome offset and scale differences between channels, and can thus sense eithe transcapacitance or self-capacitance of all tablet rows or columns in parallel. This parallel-sensin capability, made possible by providing one set of electronics per row or column, allows th sensing cycle to be extremely short, thus allowing fast response while still maintaining immunit to very high levels of electrical interference. The present invention comprises a position-sensing technology particularly usefu for applications where finger position information is needed, such as in computer "mouse" o trackball environments. However the position-sensing technology of the present invention has much more general application than a computer mouse, because its sensor can detect and report i one or more points are being touched. In addition, the detector can sense the pressure of die touch. According to a preferred embodiment of the present invention, referred to herein as a "finger pointer" embodiment, a position sensing system includes a position sensing transduce comprising a touch-sensitive surface disposed on a substrate, such as a printed circuit board, including a matrix of conductive lines. A first set of conductive lines runs in a first direction and i insulated from a second set of conductive lines running in a second direction generally perpendicular to die first direction. An insulating layer is disposed over the first and second set of conductive lines. The insulating layer is thin enough to promote significant capacitive coupling between a finger placed on its surface and the first and second sets of conductive lines. Sensing electronics respond to die proximity of a finger, conductive object, or an object of high dielectric constant (i.e., greater than about 5) to translate die capacitance changes o die conductors caused by object proximity into digital information which is processed to derive position and touch pressure information. Its output is a simple X, Y and pressure value of the one object on its surface. In all descriptions herein, fingers are to be considered interchangable with conductive objects and objects of high dielectric constant. Different prior art pad scan techniques have different advantages in different environments. Parallel drive/sense techniques according to die present invention allow input samples to be taken simultaneously, thus all channels are affected by the same phase of an interfering electrical signal, gready simplifying the signal processing and noise filtering. There are two drive/sense methods employed in die touch sensing technology of the present invention. According to a first and presendy preferred embodiment of the invention, the voltages on all of die X lines of the sensor matrix are simultaneously moved, while die voltages of the Y lines are held at a constant voltage, with the complete set of sampled points simultaneously giving a profile of the finger in the X dimension. Next, d e voltages on all of the Y lines of the sensor matrix are simultaneously moved, while die voltages of die X lines are held at a constant voltage to obtain a complete set of sampled points simultaneously giving a profile of the finger in the other dimension. According to a second drive/sense method, die voltages on all of die X lines of die sensor matrix are simultaneously moved in a positive direction, while the voltages of die Y lines are moved in a negative direction. Next, the voltages on all of d e X lines of the sensor matrix are simultaneously moved in a negative direction, while the voltages of the Y lines are moved in a positive direction. This technique doubles die effect of any transcapacitance between the two dimensions, or conversely, halves the effect of any parasitic capacitance to ground. In both methods, die capacitive information from the sensing process provides a profile of the proximity of d e finger to die sensor in each dimension. Both embodiments then take these profiles and derive a digital value representing die centroid for X and Y position and derive a second digital value for the Z pressure information. The digital information may be direcdy used by a host computer. The position sensor of these embodiments can only report die position of one object on its sensor surface. If more dian one object is present, the position sensor of tiiis embodiment computes die centroid position of the combined set of objects. However, unlike prior art, because the entire pad is being profiled, enough information is available to discern simple multi-finger gestures to allow for a more powerful user interface. According to another aspect of die present invention, several power reduction techniques which can shut down d e circuit between measurements have been integrated into the system. This is possible because the parallel measurement technique according to the present invention is so much faster than prior art techniques. According to a further aspect of d e invention, a variety of noise reductio techniques are integrated into the system. According to yet another aspect of the present invention, a capacitance measurement technique which is easier to calibrate and implement is employed. Brief Description Of The Drawings Figure 1 is an overall block diagram of the capacitive position sensing system o die present invention. Figure 2a is a top view of an object position sensor transducer according to a presendy preferred embodiment of the invention showing die object position sensor surface laye including a top conductive trace layer and conductive pads connected to a bottom trace layer. Figure 2b is a bottom view of the object position sensor transducer of FIG. 2a showing the bottom conductive trace layer. Figure 2c is a composite view of the object position sensor transducer of FIGS. 2a and 2b showing both the top and bottom conductive trace layers. Figure 2d is a cross-sectional view of the object position sensor transducer of FIGS. 2a-2c. Figure 3 is a block diagram of sensor decoding electronics which may be used witii the sensor transducer in accordance with a preferred embodiment of die present invention. Figure 4a is a simplified schematic diagram of a charge integrator circuit which may be used in the present invention. Figure 4b is an illustrative schematic diagram of die charge integrator circuit of FIG. 4a. Figure 5 is a timing diagram of die operation of charge integrator circuit of FIGS. 4a and 4b. Figure 6 is a schematic diagram of an illustrative filter and sample/hold circuit for use in die present invention. Figure 7 is a more detailed block diagram of a presendy preferred arrangement of A/D converters for use in die present invention. Figure 8 is a block diagram of an illustrative arithmetic unit which may be used in die present invention. Figure 9 is a block diagram of a calibration unit which may be used with die arithmetic unit of Figure 8. Figure 10 is a schematic diagram of a bias voltage generating circuit useful in die present invention. Detailed Description Of A Preferred Embodiment This application is a continuation-in-part of co-pending application serial No. 08/115,743, filed August 31, 1993, now United States Patent No. , which is a continuation-in-part of co-pending application serial No. 07/895,934, filed June 8, 1992. The present invention continues the approach disclosed in die parent applications and provides more unique features not previously available. These improvements provide a more easily integrated solution, increased sensitivity, and greater noise rejection, increased data acquisition rate and decreased power consumption. Additionally, the present invention allows for continuous self calibration to subtract out die effects of environmental changes. Those of ordinary skill in die art will realize that die following description of die present invention is illustrative only and not in any way limiting. Other embodiments of die invention will readily suggest tiiemselves to such skilled persons. The present invention brings together in combination a number of unique features which allow for new applications not before possible. Because the object position sensor of die present invention has very low power requirements, it is beneficial for use in battery operated or low power applications such as lap top or portable computers . It is also a very low cost solution, has no moving parts (and is therefore virtually maintenance free), and uses die existing printed circuit board traces for sensors. The sensing technology of the present invention can be integrated into a computer motiierboard to even further lower its cost in computer applications. Similarly, in otiier applications die sensor can be part of an already existent circuit board. Because of its small size and low profile, the sensor technology of die present invention is useful in lap top or portable applications where volume is an important consideration. The sensor technology of the present invention requires circuit board space for only a single sensor interface chip that can interface direcdy to a microprocessor, plus the area needed on die printed circuit board for sensing. Referring first to FIG. 1, a simplified block diagram of die capacitive position sensing system 10 of the present invention is presented. Capacitive position sensing system 10 can accurately determine d e position of a finger 12 or other conductive object proximate to or touching a sensing plane 14. The capacitance of a plurality of conductive lines running in a first direction (e.g., "X") is sensed by X input processing circuitry 16 and die capacitance of a plurality of conductive lines running in a second direction (e.g., "Y") is sensed by Y input processing circuitry 18. The sensed capacitance values are digitized in botii X input processing circuitry 16 and Y input processing circuitry 18. The outputs of X input processing circuitry 16 and Y input processing circuitry 18 are presented to aritiimetic unit 20, which uses the digital information to derive digital information representing the position and pressure of d e finger 12 or otiier conductive object relative to die sensing plane 14. The sensor material can be anything that allows creation of a conductive X/Y matrix of pads. This includes not only standard PC boards, but also includes but is not limited to flexible PC boards, conductive elastomer materials, silk-screened conductive lines, and piezo- electric Kynar plastic materials. This renders it useful as well in any portable equipment application or in human interface where the sensor needs to be molded to fit within the hand. The sensor can be conformed to any diree dimensional surface. Copper can be plated in two layers on most any surface contour producing d e sensor. This will allow d e sensor to be adapted to d e best ergonomic form needed for any particular application. This coupled with the "light-touch" feature will make it effortless to use in many applications. The sensor can also be used in an indirect manner, i.e it can have an insulating foam material covered by a conductive layer over the touch sensing surface and be used to detect any object (not just conductive) tiiat presses against it's surface. Small sensor areas are practical, i.e., a presendy conceived embodiment takes about 1.5"x 1.5" of area, however those of ordinary skill in the art will recognize that die area is scaleable for different applications. The matrix area is scalable by either varying the matrix trace spacing or by varying the number of traces. Large sensor areas are practical where more information is needed. Besides simple X and Y position information, the sensor technology of d e present invention also provides finger pressure information. This additional dimension of information may be used by programs to control special features such as "brush-width" modes in Paint programs, special menu accesses, etc., allowing provision of a more natural sensory input to computers. It has also been found useful for implementing "mouse click and drag" modes and for simple input gestures. The user will not even have to touch the surface to generate die minimum reaction. This feature can gready minimize user strain and allow for more flexible use. The sense system of the present invention depends on a transducer device capable of providing position and pressure information regarding die object contacting the transducer. Referring now to FIGS. 2a-2d, top, bottom, composite, and cross-sectional views, respectively, are shown of a presendy-preferred sensing plane 14 comprising a touch sensor array 22 for use in the present invention. Since capacitance is exploited by tiiis embodiment of die present invention, the surface of touch sensor array 22 is designed to maximize d e capacitive coupling to a finger or other conductive object A presendy preferred touch sensor array 22 according to the present invention comprises a substrate 24 including a set of first conductive traces 26 disposed on a top surface 28 thereof and run in a first direction to comprise row positions of the array. A second set of conductive traces 30 are disposed on a bottom surface 32 thereof and run in a second direction preferably orthogonal to d e first direction to form the column positions of die array. The top and bottom conductive traces 26 and 30 are alternately in contact with periodic sense pads 34 comprising enlarged areas, shown as diamonds in FIGS. 2a-2c. While sense pads 34 are shown as diamonds in FIGS. 2a-2c, any shape, such as circles, which allows diem to be closely packed is equivalent for purposes of this invention. As an arbitrary convention herein, the first conductive traces 26 will be referred to as being oriented in die "X" or "row" direction and may be referred to herein sometimes as "X lines" and die second conductive traces 30 will be referred to as being oriented in the "Y" or "column" direction and may be referred to herein sometimes as "Y lines". The number and spacing of these sense pads 34 depends upon die resolution desired. For example, in an actual embodiment constructed according to die principles of the present invention, a 0.10 inch center-to-center diamond-shaped pattern of conductive pads disposed along a matrix of 15 rows and 15 columns of conductors is employed. Every other sense pad 34 in each direction in the pad pattern is connected to conductive traces on die top and bottom surfaces 28 and 32, respectively of substrate 24. Substrate 24 may be a printed circuit board, a flexible circuit board or any of number of available circuit interconnect technology structures. Its thickness is unimportant as lon as contact may be made therethrough from die bottom conductive traces 30 to tiieir sense pads 3 on the top surface 28. The printed circuit board comprising substrate 24 can be constructed usin standard industry techniques. Board thickness is not important Connections from the conductiv pads 34 to die bottom traces 30 may be made employing standard plated-dirough hole technique well known in the printed circuit board art In an alternate embodiment of the present invention, the substrate material 24 ma have a thickness on die order of 0.005 to 0.010 inches. Then the diamonds on die upper surfac 28 and d e plated through holes that connect to the lower surface traces 30, can be omitted, furthe reducing die cost of the system. An insulating layer 36 is disposed over die sense pads 34 on top surface 28 t insulate a human finger or other object therefrom. Insulating layer 36 is preferably a thin laye (i.e., approximately 5 mils) to keep capacitive coupling large and may comprise a material, such a mylar, chosen for its protective and ergonomic characteristics. The term "significant capacitiv coupling" as used herein shall mean capacitive coupling having a magnitude greater than about 0. pF. There are two different capacitive effects taking place when a finger approaches the touch sensor array 22. The first capacitive effect is trans-capacitance, or coupling between sens pads 34, and d e second capacitive effect is self-capacitance, or coupling to virtual ground Sensing circuitry is coupled to d e sensor array 22 of the present invention and responds t changes in either or both of these capacitances. This is important because d e relative sizes of the two capacitances change gready depending on the user environment. The ability of the presen invention to detect changes in both self capacitance and trans-capacitance results in a very versatile system having a wide range of applications. According to die preferred embodiment of the invention, a position sensor system including touch sensor array 22 and associated position detection circuitry will detect a finger position on a matrix of printed circuit board traces via d e capacitive effect of finger proximity t die sensor array 22. The position sensor system will report the X, Y position of a finger placed near the sensor array 22 to much finer resolution than die spacing between die row and colum traces 26 and 30. The position sensor according to this embodiment of the invention will als report a Z value proportional to the oudine of that finger and hence indicative of die pressure widi which the finger contacts die surface of insulating layer 36 over the sensing array 22. According to the presendy preferred embodiment of die invention, a very sensitive, light-touch detector circuit may be provided using adaptive analog and digital VLSI techniques. The circuit of the present invention is very robust and calibrates out process and systematic errors. The detector circuit of the present invention will process the capacitive input information and provide digital information which may be presented direcdy to a microprocessor. According to this embodiment of the invention, sensing circuitry is contained on a single sensor processor integrated circuit chip. The sensor processor chip can have any number of X and Y "matrix" inputs. The number of X and Y inputs does not have to be equal. The Integrated circuit has a digital bus as output. In die illustrative example disclosed in FIGS. 2a-2d herein, the sensor array has 15 traces in both die X and Y directions. The sensor processor chip thus has 15 X inputs and 15 Y inputs. An actual embodiment constructed according to the principles of die present invention employed 18 traces in the X direction and 24 traces in die Y direction. Those of ordinary skill in die art will recognize that d e size of d e sensing matrix which may be employed in the present invention is arbitrary and will be dictated largely by design choice. The X and Y matrix nodes are driven and sensed in parallel, witii die capacitive information from each line indicating how close a finger is to that node. The scanned information provides a profile of die finger proximity in each dimension. According to tiiis aspect of the present invention, the profile centroid is derived in both die X and Y directions and is the position in tiiat dimension. The profile curve of proximity is also integrated to provide die Z information. There are two drive and sense methods employed in the touch sensing technology of die present invention. According to a first and presendy preferred embodiment of die invention, the voltages on all of die X lines of the sensor matrix are simultaneously moved, while the voltages of d e Y lines are held at a constant voltage. Next, the voltages on all of die Y lines of die sensor matrix are simultaneously moved, while the voltages of the X lines are held at a constant voltage. This scanning method accentuates the measurement of capacitance to virtual ground provided by die finger. Those of ordinary skill in the art will recognize that order of tiiese two steps is somewhat arbitrary and may be reversed. According to a second drive/sense metiiod, the voltages on all of die X lines of the sensor matrix are simultaneously moved in a positive direction, while die voltages of die Y lines are moved in a negative direction. Next, the voltages on all of the X lines of the sensor matrix are simultaneously moved in a negative direction, while the voltages of die Y lines are moved in a positive direction. This second drive/sense method accentuates transcapacitance and de- emphasizes virtual ground capacitance. As with the first drive/sense method, those of ordinary skill in the art will recognize that order of these two steps is somewhat arbitrary and may be reversed. Referring now to FIG. 3, a block diagram of the presently preferred sensin circuitry 40 for use according to die present invention is presented. This block diagram, and di accompanying disclosure, relates to the sensing circuitry in one dimension (X) only, and include the X input processing circuitry 16 of FIG. 1. Those of ordinary skill in the art will appreciate th an identical circuit would be used for sensing the opposite (Y) dimension and would include the input processing circuitry 18 of FIG. 1. Such skilled persons will further note tiiat the tw dimensions do not need to be orthogonal to one anotiier. For example, tiiey can be radial or of an odier nature to match die contour of die touch sensor array and other needs of die system. Tho of ordinary skill in the art will recognize that the technology disclosed herein could be applied well to a one-dimensional case where only one set of conductive traces is used. The capacitance at each sensor matrix node is represented by equivalent capacitor 42-1 dirough 42-n. The capacitance of capacitors 42-1 dirough 42-n comprises the capacitance o the matrix conductors and has a characteristic background value when no object (e.g., a finger) i proximate to the sensing plane of die sensor matrix. As an object approaches the sensing plane th capacitance of capacitors 42-1 through 42-n increases in proportion to the size and proximity of th object. According to die present invention, die capacitance at each sensor matrix node measured simultaneously using charge integrator circuits 44-1 through 44-n. Charge-integrato circuits 44-1 through 44-n serve to inject charge into the capacitances 42-1 through 42- respectively, and to develop an output voltage proportional to die capacitance sensed on di corresponding X matrix line. Thus charge-integrator circuits 44-1 through 44-n are shown a bidirectional amplifier symbols. Each charge-integrator circuit 44-1 through 44-n is supplied wi an operating bias voltage by bias-voltage generating circuit 46. As used herein, the phrase "proportional to the capacitance" means that die voltag signal generated is a monotonic function of d e sensed capacitance. In d e embodiment describe herein, the voltage is direcdy and linearly proportional to die capacitance sensed. Those o ordinary skill in the art will recognize that other monotonic functions, including but not limited t inverse proportionality, and non-linear proportionality such as logarithmic or exponenti functions, could be employed in die present invention widiout departing from die principle disclosed herein. In addition current-sensing as well as voltage-sensing techniques could b employed. According to a presently preferred drive/sense method used in the presen invention, die capacitance measurements are performed simultaneously across all inputs in one dimension to overcome a problem which is inherent in all prior art approaches that scan individual inputs. The problem with d e prior-art approach is that it is sensitive to high frequency and large amplitude noise (large dv/dt noise) that is coupled to the circuit via the touching object Such noise may distort the finger profile because of noise appearing in a later scan cycle but not an earlier one, due to a change in die noise level. The present invention overcomes this problem by "taking a snapshot" of all inputs simultaneously in X and then Y directions (or visa versa). Because the injected noise is proportional to the finger signal strength across all inputs, it is therefore symmetric around die finger centroid. Because it is symmetric around die finger centroid it does not affect die finger position. Additionally, die charge amplifier performs a differential measuring function to further reject common-mode noise. Because of the nature of the charge integrator circuits 44-1 dirough 44-n, tiieir outputs will be changing over time and will have die desired voltage output for only a short time. As presently preferred, filter circuits 48-1 dirough 48-n are implemented as sample and hold switched capacitor filters. The desired voltage is captured by die filter circuits 48-1 through 48-n. As controlled by control circuitry, 56, die filter circuits 48-1 through 48-n will filter out any high frequency noise from the sensed signal. This is accomplished by choosing the capacitor for the filter to be much larger than die output capacitiance of charge integrator circuits 44- 1 dirough 44-n. In addition, tiiose of ordinary skill in the art will recognize that die switched capacitor filter circuits 48-1 through 48-n will capture the desired voltages and store them. According to die present invention, d e capacitance information obtained in voltage form from die capacitance measurements is digitized and processed in digital format Accordingly, the voltages stored by filter circuits 48-1 through 48-n are stored in sample/hold circuits 50-1 dirough 50-n so that die remainder of the circuitry processes input data taken at die same time. Sample/hold circuits 50-1 through 50-n may be configured as conventional sample/hold circuits as is well known in the art. The sampled analog voltages at the outputs of sample/hold circuits 50-1 dirough 50-n are digitized by analog-to-digital (A/D) converters 52. As presendy preferred, A/D converters 52 resolve die input voltage to a 10-bit wide digital signal (a resolution of one part in 1,024), although diose of ordinary skill in die art will realize that other resolutions may be employed. A/D converters 52 may be conventional successive approximation type converters as is known in the art. Given the charge integrator circuitry employed in the present invention, die background level (no object present) of the charge integrator outputs will be about 1 volt. The ΔV resulting from die presence of a finger or other object will typically be about 0.4 volt. The voltage range of the A/D converters 52 should therefore be in the range of between about 1-2 volts. An important consideration is the minimum and maximum voltage reference points for the A/D converters (V min and V max ). It has been found tiiat noise will cause position jitter if tiiese reference voltages are fixed points. A solution to tiiis problem which is employed in the present invention is to dynamically generate d e V-^- and V ma -. reference voltages from reference capacitances 42- Vmin and 42-Vmax, sensed by charge integrator circuits 44- Vmin and 44-Vmax and processed by filter circuits 48-Vmin and 48-Vmax and stored in sample/hold circuits 50- Vmin and 50-Vmax . In this manner, any common mode noise present when die signals are sampled from the sensor array will also be present in the V.^-, and V max reference voltage values and will tend to cancel. Those of ordinary skill in the art will realize that reference capacitances 44- Vmin and 44-Vmax may eitiier be discrete capacitors or extra traces in die sensor array. According to die present invention, the V^ reference voltage is generated from a capacitor having a value equal to the lowest capacitance expected to be encountered in die sensor array with no object present (about 12pF assuming a 2 inch square sensor array). The V max reference voltage is generated from a capacitor having a value equal to die largest capacitance expected to be encountered in die sensor array with an object present (about 16pF assuming a 2 inch square sensor array). The outputs of A/D converters 52 provide inputs to arithmetic unit 20. As will be more fully disclosed witii reference to FIG. 8, the function of arithmetic unit 20 is to compute the weighted average of the signals on the individual sense lines in both die X and Y directions in the touch sensor array 22. Thus, aritiimetic unit 20 is shared by die X input processing circuitry 16 and die Y input processing circuitry 18 as shown in FIG. 1. Control circuitry 56 of FIG. 3 orchestrates the operation of die remainder of die circuitry. Because the system is discretely sampled and pipelined in its operation, control circuitry 56 is present to manage the signal flow. The functions performed by control circuitry 56 may be conventionally developed via what is commonly known in the art as a state machine or microcontroller. The structure and operation of the individual blocks of FIG. 3 will now be disclosed. Referring now to FIGS. 4a, 4b, and 5, a typical charge integrator circuit will be described. Charge integrator circuit 44 is shown as a simplified schematic diagram in FIG. 4a and as an illustrative schematic diagram in FIG. 4b. The timing of the operation of charge integrator circuit 44 is shown in FIG. 5. These timing signals are provided by die controller block 56. Charge integrator circuit 44 is based on die fundamental physical phenomena of using a current to charge a capacitor. If the capacitor is charged for a constant time by a constant current, then a voltage will be produced on die capacitor which is inversely proportional to the capacitance. The capacitance to be charged is the sensor matrix line capacitance 42 in parallel with an internal capacitor. This internal capacitor will contain the voltage of interest. Referring now to FIG. 4a, a simplified schematic diagram of an illustrative charge integrator circuit 44 is shown. A charge integrator circuit input node 60 is connected to one of the X (or Y) lines of the sensor matrix. A first shorting switch 62 is connected between die charge integrator circuit input node 60 and V DD , die positive supply rail. A second shorting switch 64 is connected between the charge integrator circuit input node 60 and ground, die negative supply rail. A positive constant current source 66 is connected to V D D, the positive supply rail and to the charge integrator circuit input node 60 and dirough a first current source switch 68. A negative constant current source 70 is connected to ground and to the charge integrator circuit input node 60 and dirough a second current source switch 72. It is obvious that other high and low voltage rails could be used in place of V DD and ground. A first internal capacitor 74 is connected between V DD and output node 76 of charge integrator circuit 44. A positive voltage storage switch 78 is connected between output node 76 and input node 60. A second internal capacitor 80 has one of its plates connected to ground dirough a switch 82 and to output node 76 of charge integrator circuit 44 through a switch 84, and die other one of its plates connected to input node 60 dirough a negative voltage storage switch 86 and to V DD through a switch 88. The capacitance of first and second internal capacitances 74 and 80 should be a small fraction (i.e., about 10%) of d e capacitance of the individual sensor matrix lines. In a typical embodiment die sensor matrix line capacitance will be about lOpF and die capacitance of capacitors 74 and 80 should be about lpF. According to the presendy preferred embodiment of die invention, the approach used is a differential measurement for added noise immunity, die benefit of which is tiiat any low frequency common mode noise gets subtracted out. For the following discussion, it is to be assumed that all switches are open unless they are noted as closed. First die sensor matrix line is momentarily shorted to V DD through switch 62, switch 78 is closed connecting capacitor 74 in parallel with the capacitance of the sensor line. Then the parallel capacitor combination is discharged witii a constant current from current source 70 through switch 72 for a fixed time period. At d e end of the fixed time period, switch 78 is opened, tiius storing the voltage on die sensor matrix line on capacitor 74. The sensor line is then momentarily shorted to ground through switch 64, and switches 82 and 86 are closed to place capacitor 80 in parallel with the capacitance of die sensor line. Switch 68 is closed and the parallel capacitor combination is charged with a constant current from current source 66 for a fixed time period equal to the fixed time period of die first cycle. At the end of die fixed time period, switch 86 is opened, tiius storing die voltage on die sensor matrix line on capacitor 80. The first and second measured voltages are then averaged. This is accomplished by opening switch 82 and closing switches 88 and 84, which places capacitor 80 in parallel with capacitor 74. Because capacitors 74 and 80 have die same capacitance, die resulting voltage across diem is equal to the average of the voltages across each individually. This final result is the value that is then passed on to die appropriate one of filter circuits 48-1 through 48-n. The low frequency noise, notably 50/60 Hz and their harmonics, behaves as a DC current component tiiat adds in one measurement and subtracts in die other. When the two results are added together that noise component averages to zero. The amount of noise rejection is a function of how quickly in succession the two opposing charge-up and charge-down cycles are performed as will be disclosed herein. One of the reasons for die choice of tiiis charge integrator circuit is that it allows measurements to be taken quickly. Referring now to FIG. 4b, a more complete schematic diagram of an illustrative embodiment of charge integrator circuit 44 of the simplified diagram of FIG. 4a is shown. Input node 60 is shown connected to V D D and ground through pass gates 90 and 92, which replace switches 62 and 64 of FIG. 4a. Pass gate 90 is controlled by a signal ResetUp presented to its control input and pass gate 92 is controlled by a signal ResetDn presented to its control input Those of ordinary skill in the art will recognize that pass gates 90 and 92, as well as all of die otiier pass gates which are represented by die same symbol in FIG. 4b may be conventional CMOS pass gates as are known in the art. The convention used herein is that die pass gate will be off when its control input is held low and will be on and present a low impedance connection when its control input is held high. P-Channel MOS transistors 94 and 96 are configured as a current mirror. P- Channel MOS transistor 94 serves as the current source 66 and pass gate 98 serves as switch 68 of FIG. 4a. The control input of pass gate 98 is controlled by a signal StepUp. N-Channel MOS transistors 100 and 102 are also configured as a current mirror. N-Channel MOS transistor 100 serves as the current source 70 and pass gate 104 serves as switch 72 of FIG. 4a. The control input of pass gate 104 is controlled by a signal StepDn. P-Channel MOS transistor 106 and N-Channel MOS transistor 108 are placed in series witii P-Channel MOS current mirror transistor 96 and N-Channel MOS current mirror transistor 102. The control gate of P-Channel MOS transistor 106 is driven by an enable signal EN, which turns on P-Channel MOS transistor 106 to energize die current mirrors. This device is used as a power conservation device so diat the charge integrator circuit 44 may be turned off to conserve power when it is not in use. N-Channel MOS transistor 108 has its gate driven by a reference voltage Vbias, which sets the current through current mirror transistors 96 and 108. The voltage Vbias is set by a servo feedback circuit as will be disclosed in more detail witii reference to FIG. 10. Those of ordinary skill in the art will appreciate that this embodiment allows calibration to occur in real time (via long time constant feedback) thereby zeroing out any long term effects due to sensor environmental changes. In a current embodiment of die invention, Vbias is common for all charge integrator circuits 44-1 through 44-n and 44-Vmax and 44- Vmin. Note that proper sizing of MOS transistors 102 and 108 may provide temperature compensation. This is accomplished by taking advantage of die fact tiiat d e threshold of N- Channel MOS transistor 108 reduces witii temperature while the mobility of both N-Channel MOS transistors 102 and 108 reduce with temperature. The threshold reduction has die effect of increasing die current while die mobility reduction has die effect of decreasing die current By proper device sizing these effects can cancel each other out over a significant part of the operating range. Capacitor 74 has one plate connected to V DD and die otiier plate connected to die output node 76 and to die input node 60 dirough pass gate 110, shown as switch 78 in FIG. 4a. The control input of pass gate 110 is driven by the control signal SUp. One plate of capacitor 80 is connected to input node 60 dirough pass gate 112 (switch 86 in FIG. 4a) and to VDD dirough pass gate 114 (switch 82 in FIG.4a). The control input of pass gate 112 is driven by die control signal SDn and die control input of pass gate 114 is driven by die control signal ChUp. The other plate of capacitor 80 is connected to ground through N-Channel MOS transistor 116 (switch 82 in FIG. 4a) and to output node 76 through pass gate 118 (switch 84 in FIG. 4a). The control input of pass gate 118 is driven by control signal Share. Referring now to FIGS. 4a, 4b and die timing diagram of FIG. 5, die operation of charge integrator circuit 44 during one scan cycle may be observed. First the EN (enable) control signal goes active by going to Ov. This turns on die current mirrors and energizes the charge and discharge current sources, MOS transistors 94 and 100. The ResetUp control signal is active high at this time, which shorts the input node 60 (and die sensor line to which it is connected) to V DD . The SUp control signal is also active high at tiiis time which connects capacitor 74 and die output node 76 to input node 60. This arrangement guarantees that the following discharge portion of the operating cycle always starts from a known equilibrium state. The discharge process starts after ResetUp control signal goes inactive. The StepDn control signal goes active, connecting MOS transistor 100, the discharge current source, to the input node 60 and its associated sensor line. StepDn is active for a set amount of time, and die negative constant current source discharges die combined capacitance of the sensor line and capacitor 74tivus lowering its voltage during tiiat time. StepDn is tiien turned off. A short time later d e SUp control signal goes inactive, storing the measured voltage on capacitor 74. That ends die discharge cycle. Next, die ResetDn control signal becomes active and shorts the sensor line to ground. Simultaneously die SDn and ChDn control signals become active and connect capacitor 80 between ground and the sensor line. Capacitor 80 is discharged to ground, guaranteeing that the following charge up cycle always starts from a known state. The charge up cycle starts after ResetDn control signal becomes inactive and die StepUp control signal becomes active. At tiiis point the current charging source, MOS transistor 94, is connected to the sensor line and supplies a constant surrent to charge the sensor line by increasing die voltage tiiereon. The StepUp control signal is active for a set amount of time (preferably equal to the time for the previously mentioned cycle) allowing the capacitance to charge, and dien it is turned off. The SDn control signal then goes inactive, leaving die measured voltage across capacitor 80. The averaging cycle now starts. First the voltage on capacitor 80 is level shifted. This is done by die ChDn control signal going inactive, letting one plate of the capacitor 80 float Then the ChUp control signal goes active, connecting die second plate of die capacitor to V DD . Then die Share control signal becomes active which connects the first plate of capacitor 80 to output node 76, thus placing capacitors 74 and 80 in parallel. This has die effect of averaging die voltages across die two capacitors, tiius subtracting out common-mode noise as previously described. This average voltage is also then available on output node 76. Those of ordinary skill in die art will recognize that die common-mode noise- cancelling feature inherent in the averaging of d e voltages obtained in d e discharge and charge cycles is most effective when the two cycles are performed very close together in time. According to d e present invention, the ChDn and ChUp signals should be asserted witii respect to each other within a time period much less than a quarter of the period of the noise to be cancelled in order to take advantage of tiiis feature of the present invention. According to the present invention, two different drive/sense methods have been disclosed. Those of ordinary skill in die art will readily observe tiiat die charge integrator circuit 44 disclosed widi reference to FIGS. 4a, 4b, and 5 is adaptable to operate according to eitiier scanning method disclosed herein. As is clear from an understanding of die operation of charge integrator circuit 44, its output voltage is only available for a short period of time and is subject to environmental noise. In order to minimize die effects of noise, a switched capacitor filter circuit 48 is used. Referring now to FIG. 6, a schematic diagram of an illustrative switched capacitor filter circuit 48 which may be used in die present invention is shown. Those of ordinary skill in die art will recognize this switched capacitor filter circuit which comprises an input node 120, a pass gate 122 having a control input driven by a Sample control signal, a capacitor 124 connected between d e output of die pass gate 126 and a fixed voltage such as ground, and an output node comprising d e common connection between the capacitor 124 and d e output of die pass gate 126. In a typical embodiment capacitor 116 will have a capacitance of about 10 pF. As will be appreciated by persons of ordinary skill in d e art, the switched capacitor filter 48 is in part a sample/hold circuit and has a filter time constant which is K times d e period of sample, where K is die ratio of capacitor 124 to die sum of capacitors 74 and 80 of d e charge integrator circuit 44 of FIGS. 4a and 4b to which it is connected. The switched capacitor filter circuit 48 further reduces noise injection in the system. In the preferred embodiment K = 10/2 = 5. Those of ordinary skill in die art will recognize diat other types of filter circuits, such as RC filters, may be employed in the present invention. Referring now to FIG. 7, a more detailed block diagram of a presendy preferred arrangement of A/D converters 52 of FIG. 3 is presented. There are fewer A/D converters than there are lines in die touch sensor array, and die inputs to d e A/D converters are multiplexed to share each of the individual A D converters among several lines in the touch sensor array. The arrangement in FIG. 7 is more efficient in d e use of integrated circuit layout area than providing individual A/D converters for each input line. In die embodiment illustrated in FIG. 7, twenty-four conductive line traces are assumed for die sensor array 10 of FIGS. 2a-2d. As shown in FIG. 7, the outputs of sample/hold circuits 50-1 through 50-24 are fed to the analog data inputs of analog multiplexer 130. Analog multiplexer 130 has six outputs, each of which drives the input of an individual A/D converter 52- 1 dirough 52-6. The internal arrangement of analog multiplexer 130 is such tiiat four different ones of die inputs are multiplexed to each of die outputs. Analog multiplexer 130 has been conceptually drawn as six internal multiplexer blocks 132-1 through 132-6. In die example shown in FIG. 7, inputs taken from sample/hold circuits 50-1 dirough 50-4 are multiplexed to the output of internal multiplexer block 132-1 which drives A/D converter 52-1. Similarly, inputs taken from sample/hold circuits 50-5 through 50-8 are multiplexed to the output of internal multiplexer block 132-2 which drives A/D converter 52-2; inputs taken from sample/hold circuits 50-9 through 50-12 are multiplexed to the output of internal multiplexer block 132-3 which drives A/D converter 52-3; inputs taken from sample/hold circuits 50-13 through 50-16 are multiplexed to the output of internal multiplexer block 132-4 which drives A/D converter 52-4; inputs taken from sample/hold circuits 50-17 through 50-20 are multiplexed to the output of internal multiplexer block 132-51 which drives A/D converter 52-5; and inputs taken from sample/hold circuits 50-21 dirough 50-24 are multiplexed to die output of internal multiplexer block 132-6 which drives A/D converter 52-6. Analog multiplexer 130 has a set of control inputs schematically represented by bus 134. In the illustrative embodiment shown in FIG. 7, each of internal multiplexers 132-1 dirough 132-6 are four-input multiplexers and tiius control bus 134 may comprise a two-bit bus for a one- of-four selection. Those of ordinary skill in the art will recognize tiiat die arrangement of FIG. 7 is merely one of a number of specific solutions to the task of A/D conversion from twenty-four channels, and that other satisfactory equivalent arrangements are possible. In a straightforward decoding scheme, multiplexers 132-1 through 132-6 will pass, in sequence, the analog voltages present on their first through fourth inputs on to die inputs of A/D converters 52-1 through 52-6 respectively. After the analog values have settled in die inputs of A/D converters 52-1 through 52-6, a CONVERT command is asserted on common A/D control line 136 to begin the A/D conversion process. When the A/D conversion process is complete, die digital value representing the input voltage is stored in registers 138-1 through 138-6. As presendy preferred, registers 138-1 dirough 138-6 may each comprise a two- word register, so tiiat one word may be read out of die registers to arithmetic unit 54 while a second word is being written into the registers in order to maximize die speed of die system. The design of such registers is conventional in the art Referring now to FIG. 8, a more detailed block diagram of the arithmetic unit 20 is presented. Those of ordinary skill in the art will appreciate tiiat aritiimetic unit 20 processes information from both the X and Y dimensions, i.e., from X input processing circuit 16 and Y input processing circuit 18 of FIG. 1. Before disclosing the structural configuration of arithmetic unit 20, it is helpful to understand die preferred mediod by which the centroid position of an object proximate to the sensor array 22 is determined according to die present invention. According to a presendy preferred embodiment of die invention, the object position in either direction may be determined by evaluating the weighted average of die capacitances measured on the individual sense line of die sensor array 10. In the following discussion, die X direction is used, but those of ordinary skill in the art will recognize that the discussion applies to die determination of the weighted average in the Y direction as well. As is well known, die weighted average may be determined as follows: n ∑ i x ΔCi X position = ~ [Eq. 1] ∑ ΔC/ i»0 where ΔC • = C, - CO,. C • is the capacitance presendy being measured on die ith trace and CO, is the value measured on tiiat same trace at some past time when no object was present. In terms of these past and present capacitance measurements, die position can be expressed as: n ∑ ix (Cι - C0i) Xposition = -*- -- [Eq. 2] ∑ (Ci - CO/) i «0 Using the distributive property of multiplication over addition, tiiis expression is seen to be equivalent to: n n - ∑ (ι χ C0ι) + ∑ (ι ) Xposition = —- '^— u -^ [Eq. 3] - ∑ (CO,) + ∑ (CO where the negative terms in both the numerator and denominator are offsets and represent th background value of the capacitances with no object present. If the term O N is used to represen the numerator offset and die term O D is used to represent the denominator offset, Eq. 3 may be re written as: n - ON + ∑ (i x Cι) 1=0 Xposition = - [Eq. 4] - OD + ∑ (C0 i=0 Referring now to FIG. 8, it may be seen tiiat aritiimetic unit 20 includes numerator and denominator accumulators 150 and 152 and Y numerator and denominato accumulators 154 and 156. The source of operand data for X numerator and denominato accumulators 150 and 152 and Y numerator and denominator accumulators 154 and 156 are th registers 138-1 through 138-6 in each (X and Y) direction of the sensor array 22 of FIG. 1. Th X and Y denominator accumulators 152 and 156 sum up the digital results from die A/ conversions. The X and Y numerator accumulators 150 and 154 compute die weighted sum of di input data rather tiian the straight sum. Accumulators 150, 152, 154, and 156 may be configure as hardware elements or as software running on a microprocessor as will be readily understood b those of ordinary skill in the art. As may be seen from an examination of FIG. 8, numerator accumulators 150 an 154 compute the expression of Eq. 4: n Σ < χ Q i-0 and denominator accumulators 152 and 156 compute die expression of Eq.4: n The contents of X and Y numerator and denominator offset registers 158, 160, 162, and 164 are subtracted from die results stored in the accumulators 150, 152, 154, and 156 in adders 166, 168, 170, and 172. Adder 166 subtracts the offset O NX stored in X numerator offset register 158. Adder 168 subtracts the offset O DX stored in X denominator offset register 160. Adder 170 subtracts the offset Owy stored in X numerator offset register 162. Adder 172 subtracts the offset Orjγ stored in Y denominator offset register 164. The numerator denominator pairs are divided by division blocks 174 and 176 to produce the X and Y position data, and die X and Y denominator pair is used by block 178 to produce Z axis (pressure) data. The function performed by block 178 will be disclosed later herein. The offsets O D χ, O NX , O D γ, and O NY are sampled from the accumulator contents when directed by calibration unit 180. Persons of ordinary skill in the art will readily appreciate tiiat the architecture of the system of die present invention may be distributed in a number of ways, several of which involve the availability of a microprocessor, whether it be in a host computer to which the system of the present invention is connected or somewhere between die integrated circuit described herein and a host computer. Embodiments of the present invention are contemplated wherein d e accumulated numerator and denominator values representing the summation terms are delivered to such a microprocessor along with die O N and O D offset values for processing, or where all processing is accomplished by a programmed microprocessor as is known in the art Initially, the numerator and denominator accumulators 150, 152, 154, and 156 are set to zero during system startup. If the multiplexed A D converters as shown in FIG. 7 are employed, die digitized voltage data in die first word of register 138-1 (representing die voltage at die output of sample/hold circuit 50-1) is added to d e sum in die accumulator and d e result stored in d e accumulator. In succession, the digitized voltage values stored in the first word of registers 138-2 through 138-6 (representing the voltage at the outputs of sample/hold circuits 50-5, 50-9, 50-17, and 50-21, respectively) are added to the sums in the accumulators and die results stored in the accumulators. As previously mentioned, A/D converters 52-1 dirough 52-6 may at this time be converting the voltages present at the outputs of sample/hold circuits 50-2, 50-6, 50-10, 50-14, 50-18, and 50-22 and storing the digitized values in the second words of registers 138-1 through 138-6 respectively. Next, in succession, the digitized voltage values stored in die second words of registers 138-1 through 138-6 (representing the voltage at the outputs of-sample hold circuits 50- 2, 50-6, 50-10, 50-14, 50-18, and 50-22, respectively) are added to the sum in the accumulator and die result stored in the accumulator. Next, in succession, the digitized voltage values stored in the first words o registers 138-1 through 138-6 (representing die voltage at the outputs of sample/hold circuits 50- 50-7, 50-11, 50-15, 50-19, and 50-23, respectively) are added to die sum in die accumulator an die result stored in d e accumulator, followed by digitized voltage values stored in die secon words of registers 138-1 through 138-6 (representing the voltage at die outputs of sample/hol circuits 50-4, 50-8, 50-12, 50-16, 50-20, and 50-24, respectively). At this point in time, the accumulators hold the sums of all of die individua digitized voltage values. The digital values stored in die O N and O D offset registers 158 and 16 are now respectively subtracted from the values stored in the numerator and denominato accumulators. The division operation in dividers 174 and 176 tiien completes the weighte average computation. The division operation may also be performed by an external microprocesso which can fetch the values stored in the accumulators or perform die accumulations itself. As th O N and O D offset values are presently derived by an external microprocessor, die additiona processing overhead presented to such external microprocessor by this division operation i minimal. Alternately, a dedicated microprocessor may be included on chip to handle thes processing tasks without departing from die invention disclosed herein. The above disclosed processing takes place within about 1 millisecond and may b repeatedly performed. Current mouse standards update position information 40 times per second and thus the apparatus of the present invention may easily be operated at tiiis repetition rate. Because of the nature of the method employed in die present invention, a opportunity exists to provide additional noise immunity without requiring additional hardware i die system of die present invention. While it is apparent that after the above-disclosed sequenc has been performed, die accumulators may be cleared and die process repeated, die values ma also be allowed to remain in the accumulators. If this is done, an averaging function may b implemented to further filter out noise. According to tiiis aspect of the invention, a number o samples are taken and run through the accumulators widiout clearing them at die end of di processing sequence. As presently preferred, twenty-five samples are processed before a singl division result is taken for use by die system, tiius gready reducing the effects of transient syste noise spikes. Those of ordinary skill in the art will recognize that the number of samples take prior to clearing the accumulators is a matter of design choice dictated by factors such as dat acquisition rates, data processing rates etc. The system of the present invention is adaptable to changing conditions, such as component aging, changing capacitance due to humidity, and contamination of the touch surface, etc. In addition, die present invention effectively minimizes ambient noise. According to die present invention, tiiese effects are taken into consideration in three ways. First the offset values O N and O D are dynamically updated to accommodate changing conditions. Second, a servo- feedback circuit is provided to determine the bias voltage used to set the bias of the charge- integrator circuits 44-1 through 44-n. Third, as previously disclosed herein, the reference voltage points for V max and V-.^ of die A D converters are also dynamically altered to increase the signal to noise margin. Referring now to FIG. 9, a block diagram of a calibration unit 150 which may be used with the arithmetic unit of FIG. 8 is presented. The calibration unit 150 executes an algoridim to establish the numerator and denominator offset values by attempting to determine when no finger or other conductive object is proximate to the touch sensor array 22. As previously disclosed, the O N and Op offset values represent the baseline values of the array capacitances with no object present These values are also updated according to the present invention since baseline levels which are too low or too high have the effect of shifting the apparent position of the object depending on die sign of die error. These values are established by selection of the values read when no object is present at die sensor array 22. Since tiiere is no external way to "know" when no object is present at sensor array 22, an algorithm according to anodier aspect of die present invention is used to establish and dynamically update tiiese offset values. When the calibration unit sees a Z value which appears typical of the Z values when no finger is present, it instructs d e offset registers (158, 160, 162, and 164 of FIG. 8) to reload from d e current values of the accumulators. According to a presendy preferred embodiment of die invention, the decision to update die offset values is based on die behavior of die sensor array 22 in only one of die X or Y directions, but when die decision is made all four offsets (Ow,χ, Opx, ONY, and O DY ) are updated. In other embodiments of the invention, the decision to update may be individually made for each direction according to die criteria set forth herein. The calibration algoridim works by monitoring changes in a selected one of die denominator accumulator values. According to the present invention, it has been observed that die sensitivity to changes in capacitance of one of the sets of conductive lines in die touch sensor array 22 is greater than the sensitivity to changes in capacitance of the other one of the sets of conductive lines in die touch sensor array 22. Experience suggests that the set of conductive lines having the greater sensitivity to capacitance changes is die one which is physically located above the conductive lines in the otiier direction and therefore closest to the touch surface of the sensor array 22. The upper set of conductive lines tends to partially shield die lower set of conductive lines from capacitive changes occurring above the surface of the sensor array 22. The finger pressure is obtained by summing the capacitances measured on die sense lines. This value is already present in die denominator accumulator after subtracting the offset O D . A finger is present if the pressure exceeds a suitable tiireshold value. This threshold may be chosen experimentally and is a function of surface material and circuit timing. The tiireshold may be adjusted to suit the tastes of die individual user. The pressure reported by the device is a simple function f(Xn YD) °f me denominators for die X and Y directions as implemented in block 178 of FIG. 8. Possible functions include choosing one preferred denominator value, or summing the denominators. In a presently preferred embodiment, die smaller of the two denominators is chosen. This choice has the desirable effect of causing the pressure to go below the threshold if the finger moves slightiy off the edge of die pad, where die X sensors are producing valid data, but the Y sensors are not, or vise versa. This acts as an electronic bezel which can take the place of a mechanical bezel at the periphery of the sensor area. In the example of FIG. 8, the Y denominator is chosen for monitoring because it is the most sensitive. The chosen denominator is referred to as Z for die purposes of the calibration algorithm. The current saved offset value for this denominator is referred to as O^ The goal of the calibration algorithm is to track gradual variations in the resting Z level while making sure not to calibrate to the finger, nor to calibrate to instantaneous spikes arising from noise. As will be apparent to those of ordinary skill in the art from the following disclosure, die calibration algorithm could be implemented in digital or analog hardware, or in software. In a current embodiment actually tested by die inventors, it is implemented in software. As Z values arrive in the calibration unit, they are passed through filter 182. History buffer 184, which operates in conjunction with filter 182, keeps a "running average" of recent Z values. When a new Z value arrives, the current running average F z is updated according to the formula: new F z =α(old F z ) + (l- α)Z where α is a constant factor between 0 and 1 and typically close to 1 and Z is the current Z value. In die preferred embodiment, alpha is approximately 0.95. The intention is for FZ to change slowly enough to follow gradual variations, without being greatly affected by short perturbations in Z. The filter 182 receives a signal ENABLE from control unit 186. The running average F z is updated based on new Z values only when ENABLE is asserted. If ENABLE is deasserted, F z remains constant and is unaffected by current Z. The history buffer 184 records die several most recent values of FZ. In the present embodiment, die history buffer records the two previous FZ values. The history buffer might be implemented as a shift register, circular queue, or analog delay line. When the history buffer receives a REWIND signal from control unit 186, it restores the current running average F z to the oldest saved value. It is as if die filter 182 were "retroactively" disabled for an amount of time corresponding to die deptii of die history buffer. The purpose of the history buffer is to permit such retroactive disabling. The current running average F z is compared against the current Z value and d e current offset O- ζ by absolute difference units 188 and 190, and comparator 192. Absolute difference unit 188 subtracts the values Z and F z and outputs the absolute value of their difference. Absolute difference unit 1 0 subtracts the values O z and F z and outputs die absolute value of tiieir difference. Comparator 192 asserts the UPDATE signal if the output of absolute difference unit 188 is less than the output of absolute difference unit 190, i.e., if F z is closer to Z than it is to C^. The UPDATE signal will tend to be asserted when die mean value of Z shifts to a new resting level. It will tend not to be asserted when Z makes a brief excursion away from its normal resting level. The filter constant a determines die lengtii of an excursion which will be considered "brief for tiiis purpose. Subtractor unit 194 is a simple sub tractor that computes the signed difference between Z and O z . This subtractor is actually redundant with subtractor 172 in figure 8, and so may be merged with it in the actual implementation. The output C z of this subtractor is the calibrated Z value, an estimate of the finger pressure. This pressure value is compared against a positive and negative Z threshold by comparators 196 and 198. These thresholds are shown as ZJJJ and -Z jj , although they are not actually required to be equal in magnitude. If pressure signal C z is greater than Zχ H , the signal FINGER is asserted indicating the possible presence of a finger. The ZJΉ tiireshold used by the calibration unit is similar to tiiat used by the rest of the system to detect the presence of die finger, or it may have different value. In the present embodiment, the calibration Z^ is set somewhat lower than th main Z T H to ensure that die calibration unit makes a conservative choice about die presence of finger. If pressure signal C z is less than - Z τH , the signal FORCE is asserted. Since O z i meant to be equal to the resting value of Z with no finger present, and a finger can only increas the sensor capacitance and tiius the value of Z, a largely negative C z implies that the device mus have incorrecdy calibrated itself to a fmger, which has just been removed. Calibration logic 20 uses this fact to force a recalibration now that the finger is no longer present. Control logic 186 is responsible for preventing running average F z from bein influenced by Z values tiiat occur when a finger is present Output ENABLE is generally of when the FINGER signal is true, and on when the FINGER signal is false. However, when FINGER transitions from false to true, the control logic also pulses the REWIND signal. When FINGER transitions from true to false, the control logic waits a short amount of time (comparable to the deptii of the history buffer) before asserting ENABLE. Thus, the runnin average is prevented from following Z whenever a finger is present, as well as for a short time before and after die finger is present Calibration logic 200 produces signal RECAL from the outputs of the three comparators 192, 196, and 198. When RECAL is asserted, die offset registers O N and On will b reloaded from die current accumulator values. RECAL is produced from die following logic equation: RECAL = FORCE or (UPDATE and not FINGER). In addition, calibration logic 200 arranges to assert RECAL once when the system is firs initialized, possibly after a brief period to wait for the charge integrators and otiier circuits to stabilize. From the descriptions of control logic 186 and calibration logic 200, it will be apparent to those of ordinary skill in the art that these blocks can be readily configured usin conventional logic as a matter of simple and routine logic design. It should be obvious to any person of ordinary skill in the art that the calibration algoridim described is not specific to the particular system of charge integrators and accumulators of the current invention. Rather, it could be employed in any touch sensor which produces proximity or pressure data in which it is desired to maintain a calibration point reflecting the state of the sensor when no finger or spurious noise is present. Referring now to FIG. 10, a bias voltage generating circuit 46 useful in the present invention is shown in schematic diagram form. According to a presendy preferred embodiment of die invention, all of the bias transistors 108 (FIG. 4b) of charge integrator circuits 44-1 through 44-n have tiieir gates connected to a single source of bias voltage, altiiough persons of ordinary skill in die art recognize that other arrangements are possible. There are a number of ways in which to generate the bias voltage required by charge integrator circuits 44-1 through 4-n. As may be seen from an examination of FIG. 10, the bias voltage generating circuit 46 is an overdamped servo system. A reference source which approximates die current source function of a typical one of the charge integrator circuits 44-1 through 44-n includes a capacitor 204 having one of its plates grounded. The other one of its plates is connected to die V D D power supply through a first pass gate 206 and to a current source transistor 208 through a second passgate 210. A filter circuit 212, identical to the filter circuits 48-1 through 48-n and controlled by the same signal as filter circuits 48-1 through 48-n is connected to sample the voltage on capacitor 204 in die same manner that the filter-and-sample/hold circuits 48-1 through 48-n sample the voltages on the sensor conductor capacitances in the sensor array 22. The output of filter circuit 212 is fed to die non-inverting input of a weak transconductance amplifier 214, having a bias current in the range of from about 0.1-0.2μA. The inverting input of the transconductance amplifier 214 is connected to a fixed voltage of about 1 volt generated, for example, by diode 216 and resistor 218. The output of transconductance amplifier 214 is shunted by capacitor 220 and also by capacitor 222 dirough passgate 224. Capacitor 222 is chosen to be much larger than capacitor 220. In a typical embodiment of the present invention, capacitor 220 may be about 0.2pF and capacitor 222 may be about lOpF. Capacitor 222 is connected to d e gate of N-Channel MOS transistor 226, which has its drain connected to the drain and gate of P-Channel MOS transistor 228 and its source connected to die drain and gate of N-Channel MOS transistor 230. The source of P-Channel MOS transistor 228 is connected to V D D and die source of N-Channel MOS transistor 230 is connected to ground. The common drain connection of transistors 226 and 230 is die bias voltage output node. An optional passgate 232 may be connected between a fixed voltage source (e.g., about 2 volts) and capacitor 222 dirough a passgate 234. Passgate 234 may be used to initializ the bias generating circuit 200 on startup by charging capacitor 222 to the fixed voltage. During each sample period, the filter circuit 210 takes a new sample. If the ne sample differs from the previous sample, the output voltage of transconductance amplifier 211 wil change and start to charge or discharge capacitor 218 to a new voltage. Passgate 222 is switche on for a short time (i.e., about lμsec) and die voltages on capacitors 218 and 220 try to averag tiiemselves. Due to the large size difference between capacitors 218 and 220, capacitor 218 canno supply enough charge to equalize the voltage during the period when passgate 222 is open. Thi arrangement prevents large changes in bias voltage from cycle to cycle. Capacitor 202 should look as much as possible like one of the sensor arra channels and has a value equal to the background capacitance of a typical sensor line, (i.e., wit no object proximate or present capacitance component). Capacitor 202 may be formed in severa ways. Capacitor 202 may comprise an extra sensor line in a part of the sensor array, configured t approximate one of the active sensor lines but shielded from finger capacitance by a ground plane, etc. Alternately, capacitor 202 may be a capacitor formed in d e integrated circuit or connecte thereto and having a value selected to match that of a typical sensor line. In this respect, the signal source comprising capacitor 202 and filter circuit 210 is somewhat like the circuitry for generating e max and v.,, ! ,, reference voltages, in that it mimics a typical sensor line. As another alternative, one of the actual sensor lines may be employed to set d e bias voltage. The measured voltage on die two end-point sensor lines may be compared and di one having die lowest value may be selected on the theory that, if a finger or other object is proximate to the sensor array, it will not be present at sensor lines located at the opposite edges o die array. The increased sensitivity of the touch sensor system of the present invention allow for a lighter input finger touch which makes it easy for human use. Increased sensitivity als makes it easier to use other input objects, like pen styli, etc. Additionally tiiis sensitivity allow for a trade-off against a thicker protective layer, or different materials, which both allow for lower manufacturing costs. Greater noise rejection allows for greater flexibility in use and reduced sensitivit to spurious noise problems. Two techniques are employed which allow derivation of die most noise-rejection benefit Due to the drive and sense techniques employed in the present invention, the data acquisition rate has been increased by about a factor of 30 over die prior art. This offers several obvious side effects. First, for die same level of signal processing, the circuitry can be turned off most of die time and reduce power consumption by roughly a factor of 30 in the analog section of the design. Second, since more data is available, more signal processing, such as filtering, and gesture recognition, can be performed. The sensor electronic circuit employed in die present invention is very robust and calibrates out process and systematic errors. It will process the capacitive information from die sensor and provide digital information to an external device, for example, a microprocessor. Because of the unique physical features of the present invention, there are several ergonomically interesting applications that were not previously possible. Presently a Mouse or Trackball is not physically convenient to use on portable computers. The present invention provides a very convenient and easy-to-use cursor position solution that replaces those devices. In mouse-type applications, the sensor of the present invention may be placed in a convenient location, e.g., below the "space bar" key in a portable computer. When placed in tiiis location, the thumb of the user may be used as die position pointer on the sensor to control the cursor position on the computer screen. The cursor may then be moved without die need for die user's fingers to leave the keyboard. Ergonomically, tiiis is similar to the concept of the Macintosh Power Book with it's trackball, however the present invention provides a significant advantage in size over the track ball. Extensions of this basic idea are possible in that two sensors could be placed below the "space bar" key for even more feature control. The computer display with it's cursor feedback is one small example of a very general area of application where a display could be a field of lights or LED's, an LCD display, or a CRT. Examples include touch controls on laboratory equipment where present equipment uses a knob/button/touch screen combination. Because of the articulating ability of this interface, one or more of those inputs could be combined into one of d e inputs described widi respect to the present invention. Consumer Electronic Equipment (stereos, graphic equalizers, mixers) applications often utilize significant front panel surface area for slide potentiometers because variable control is needed. The present invention can provide such control in one small touch pad location. As Electronic Home Systems become more common, denser and more powerful human interface is needed. The sensor technology of the present invention permits a very dense control panel. Hand Held TV/VCR/Stereo controls could be ergonomically formed and allow for more powerful features if this sensor technology is used. The sensor of the present invention can be conformed to any surface and can b made to detect multiple touching points, making possible a more powerful joystick. The uniqu pressure detection ability of the sensor technology of the present invention is also key to thi application. Computer games, "remote" controls (hobby electronics, planes) , and machine too controls are a few examples of applications which would benefit from the sensor technology of th present invention. Musical keyboards (synthesizers, electric pianos) require velocity sensitive key which can be provided by d e pressure sensing ability of this sensor. There are also pitch bendin controls, and otiier slide switches tiiat could be replaced witii this technology. An even mo unique application comprises a musical instrument that creates notes as a function of the positio and pressure of the hands and fingers in a very articulate 3-d interface. The sensor technology of the present invention can best detect any conductin material pressing against it. By adding a compressible insulating layer covered by a layer o conductive material on top of the sensor the sensor of the present invention may also indirectl detect pressure from any object being handled, regardless of its electrical conductivity. Because of die amount of information available from this sensor it will serve ver well as an input device to virtual reality machines. It is easy to envision a construction tiiat allow position-monitoring in three dimensions and some degree of response (pressure) to actions. While embodiments and applications of tiiis invention have been shown an described, it would be apparent to those skilled in d e art that many more modifications tha mentioned above are possible witiiout departing from die inventive concepts herein. Th invention, therefore, is not to be restricted except in the spirit of the appended claims.

Claims: Claims We claim: 1. A method for providing an electrical signal representative of the position of an object in a two dimensional plane, including die steps of: providing a sensing plane including a matrix of conductors arranged as a plurality of rows and columns of spaced apart row conductive lines and column conductive lines, said sensing plane characterized by an inherent capacitance on d e various ones of said row conductive lines and column conductive lines, said capacitance varying with the proximity of an object to said row and column conductors; simultaneously developing a first set of digital signals proportional to the value of said capacitance for each of said row conductive lines when no object is located proximate to said sensing plane; simultaneously developing a second set of digital signals proportional to the value of said capacitance for each column conductive lines when no object is located proximate to said sensing plane; simultaneously developing a third set of digital signals proportional to the value of said capacitance for each of said row conductive lines when an object is located proximate to said sensing plane; simultaneously developing a fourth set of digital signals proportional to the value of said capacitance for each column conductive lines when said object is located proximate to said sensing plane; computing a first weighted average of the difference between said first set of digital signals and said third set of digital signals; computing a second weighted average of die difference between said second set of digital signals and said fourth set of digital signals. 2. The method of claim 1, wherein the steps of simultaneously developing said first second, third, and fourth sets of digital signals includes the steps of: placing a first known voltage on said capacitances; discharging said capacitances for a fixed time at a fixed current; measuring and storing a first set of resultant voltages across said capacitances; placing a second known voltage on said capacitances; charging said capacitances for said fixed time at said fixed current; measuring and storing a second set of resultant voltages across said capacitances; and averaging corresponding ones of said first and second sets of resultant voltages. 3. The method of claim 1 wherein die steps of computing said first and secon weighted averages comprises the steps of: computing a sum and a weighted sum of said first set of digital signals; computing a sum and a weighted sum of said second set of digital signals; computing a sum and a weighted sum of said third set of digital signals; computing a sum and a weighted sum of said fourth set of digital signals; computing a row numerator by subtracting said weighted sum of said first set of digit signals from said weighted sum of said third set of digital signals; computing a row denominator by subtracting said sum of said second set of digital signal from said sum of said fourth set of digital signals; dividing said row numerator by said row denominator to derive a row position digit signal representing die position of said object in a row dimension; computing a column numerator by subtracting said weighted sum of said second set o digital signals from said weighted sum of said second set of digital signals; computing a column denominator by subtracting said sum of said second set of digit signals from said sum of said second set of digital signals; and dividing said column numerator by said column denominator to derive a column positio digital signal representing the position of said object in a column dimension. 4. The method of claim 3 including the further steps of: storing said sum and said weighted sum of said first and third set of digital signals as stored sum and a stored weighted sum of said second and fourth sets of digital signals; and using said stored sum and said stored weighted sum in computing subsequent ones of sai row numerators and denominators and said column numerators and denominators using said stored stored sum and a stored weighted sum for providing an electrical signa representative of a subsequent position of said object in said two dimensional plane.

Inventors: ALLEN TIMOTHY P, GILLESPIE DAVID, MILLER ROBERT J, STEINBACH GUNTER, ALLEN, TIMOTHY, P., GILLESPIE, DAVID, MILLER, ROBERT, J., STEINBACH, GUNTER
Applicants: ALLEN TIMOTHY P, GILLESPIE DAVID, MILLER ROBERT J, STEINBACH GUNTER, SYNAPTICS INC, ALLEN, TIMOTHY, P., GILLESPIE, DAVID, MILLER, ROBERT, J., STEINBACH, GUNTER, SYNAPTICS, INC.
Application year: 1995
Main classification: G06K11
IPC codes: G06F3