Maximum likelihood decoding coherent detecting method

Abstract: Samples z n-q of a received signal are inversely modulated by candidate phases  n-q for q = 1 and 2, a reference signal η n is calculated using the following equation where α is a prediction coefficient: η n = (1 + α)z n-1 exp(-j n-1 )-αz n-2 exp(-j n-2 ). The square error between the reference signal η n and said phase-rotated sample z n is obtained as a branch metric λ(S n-1 →S n ) indicating the likelihood of transition from said state S n-1 at time point n-1 to a state S n at time point n.

Description: The present invention relates to a maximum likelihood coherent detection method which makes the maximum likelihood estimation of a reference signal for coherent detection and the maximum likelihood sequence estimation of a transmitted symbol sequence at the same time through the use of a plurality of received signal samples obtained by sampling a received phase-modulated digital signal at the symbol rate. Coherent detection and differential detection are widely used for the demodulation of phase-modulated signals. The coherent detection provides an excellent error rate performance as compared with the differential detection. To perform the coherent detection, it is necessary that the phase of the received carrier be known. To this end, the receiving side regenerates the carrier by some means, uses it as a reference signal to coherently detect the modulated phase of the received signal and decides the transmitted data. As a method of extracting an M-phase PSK reference signal, there is known a multiplying method according to which the received signal is M multiplied to remove therefrom the modulated phase, the multiplied signal is used to effect phase control of a voltage-controlled oscillator (VCO) to generate a signal of a frequency M times higher than the carrier frequency and the frequency of the signal is M demultiplied to regenerate the carrier intended to obtain. Another known method is, for example, an inverse modulation method according to which a detected data is used to inversely modulate to remove therefrom the modulated phase and than the carrier is similarly regenerated using the VCO. However, these methods are disadvantageous in that no fast carrier regeneration is possible because the carrier extracting and regenerating steps form a closed loop using the VCO. Since the regenerated carrier has a phase uncertainty of 2π/M rad., a known signal sequence (several symbols, for instance) is transmitted periodically and used to avoid the uncertainty of the phase. This coherent detection is called absolute coherent detection in the sense of detecting the absolute phase. On the other hand, it is possible to exclude the influence of the phase uncertainty by differential encoding at the sending side and differential decoding at the receiving side, but the error rate in this instance would be about twice that of the absolute coherent detection scheme. Moreover, the bit error rate performance of the absolute coherent detection is superior to that of the differential detection. In the case of four-phase PSK, the difference between the differential detection and the absolute coherent detection in the ratio E b /N o of the received energy per bit necessary for securing a 0.1% bit error rate to the noise power density is as large as 2.5 dB or so. Incidentally, there has been proposed a coherent detection scheme which does not involve the carrier regeneration quite unlike the above-mentioned methods (Literature 1: D. Divsalar and M.K. Simon, "Multiple-symbol differential detection of MPSK," IEEE Trans. Commun., vol. 38, pp. 300-308, March 1990 and Literature 2: P. Y. Kam, "Maximum-likelihood digital data sequence estimation over the Gaussian channel with unknown carrier phase," IEEE Trans. Commun., vol. COM-35, pp. 764-767, July 1987). The methods proposed in the both Literature 1 and 2 make the maximum likelihood phase sequence estimation of transmitted symbols through the use of received signal samples produced by sampling the received signal with the symbol period. A metric is calculated for each sequence composed of N symbols using N received signal samples and the sequence of the maximum metric is outputted. Hence, the carrier phase need not be regenerated. To obtain bit error rate performance close to that of ideal coherent detection, it is necessary to increase the number N of symbols for sequence estimation. Since the number of metric calculations for maximum estimation becomes M N , however, the amount of calculation increases exponentially as the number M of modulated phases and the number N of transmitted symbols increase. The document DIVSALAR et al.: "The performance of trellis-coded MDPSK with multiple symbol detection" IEEE TRANSACTIONS ON COMMUNICATIONS, vol.38, no.9, September 1990, pages 1391-1403, NEW YORK US relates to trellis-coded MDPSK with multiple symbol detection, wherein multiple trellis-coded modulation is combined with a multiple symbol differential detection scheme and the maximum likelihood metric η i for correct data phase sequence on the i-th trellis branch is calculated in accordance with an equation that does not use a weighting factor α. An object of the present invention is to provide an approximate maximum likelihood decoding coherent detection method which makes a sequential transmitted symbol sequence estimation through the use of the Viterbi algorithm but permits reduction of the amount of processing required. This object is achieved by a method as claimed in claim 1. Preferred embodiments of the invention are subject-matter of the dependent claims. BRIEF DESCRIPTION OF THE DRAWINGS Fig. 1 is a trellis diagram with M=4 and Q=1. Fig. 2 is a diagram showing an example of a transmit frame configuration. Fig. 3 is a block diagram illustrating an example of a coherent detection receiver in accordance with a first embodiment of the invention. Fig. 4 is a block diagram illustrating an example of a coherent detection receiver in accordance with a second embodiment of the invention. Fig. 5 is a graph showing error rate performance of first and second embodiments of the invention. Fig. 6 is a graph showing error rate performance of the first and second embodiments under the fading environment obtained by computer simulations. BEST MODE FOR CARRYING OUT THE INVENTION As a basis for the embodiments of the invention described later, an example will be described first using mathematical expressions. Now, consider the maximum likelihood estimation of an N-symbol phase sequence  n (n=0, 1, 2, ..., N-1) being transmitted. An M-phase PSK signal received at time (n-1)T≤t≤nT can be expressed by z(t)=(2E s /T) ½ exp{j( n +)}+w(t) where  n ={2mπ/M; m=0, 1, ..., M-1} is the modulated phase, E s the energy per symbol, T one symbol length,  the phase difference between the received signal carrier and a receiver local oscillation signal, and w(t) receiver noise. The signal z(t) is filtered and then sampled at discrete points in time n=0, 1, 2, ... for each symbol period T. Let the thus obtained sample sequence represented by {z n ; n=0, 1, 2, ..., N-1}. In Literature 1 and 2 it is described that it is the maximum likelihood sequence that maximizes the metric given by the following equation: If the maximum sequence likelihood estimation is made faithfully to Literature 1 and 2, the metric calculation has to be done M N times as referred to previously. In accordance with this example, Eq. (2) is modified first as follows: where * indicates a complex conjugate and Re[X] the real part of any complex number X. On the right side of Eq. (3) the first term is unrelated to the sequence, and hence can be omitted. In Eq. (3) the upper limit of the summation concerning q is n and its maximum value is N-1. On this account, the amount of processing required increases exponentially as the sequence grows longer. To avoid this, the upper limit of the summation concerning q is set to Q(<<N) and the path metric is defined by the following equation: By the definition like this, sequences of the maximum path metric A can be sequentially estimated for each of the M Q -states by the Viterbi algorithm with the branch metric set as follows: λ( n-1 → n )=Re{z n η n *exp(-j n )} Here, η n is given by which is the estimate of the carrier (2E s /T) ½ exp(j) in Eq. (1), and this value η n is used as a reference signal for coherent detection of the sample sequence z n . For details of the Viterbi algorithm, see literature: Imai, "Coding Theory," Institute of Electronics, Information and Communication Engineers of Japan. Fig. 1 is a trellis diagram. For brevity, suppose that M=4 and Q=1. Furthermore, survival paths SP that reach the states at time n-1 are assumed to have been determined. Each state S n at time n is represented by a combination of Q phases  n-q , where q=0, 1, 2,..... Q-1. That is to say, S n ={ n-q ; q=0, 1, 2, ..., Q-1}. The number of states is M Q in total. M paths extend to each state at time n from M Q states at time n-1 regardless of the value Q. The transition from a certain state S n-1 at time n-1 to a certain state S n at time n is represented by a combination of Q+1 phases  n-q , where q=0, 1, 2,..., Q-1, Q. The branch metric that indicates the likelihood of this transition is calculated on the basis of Eq. (5), and the branch metric thus calculated is added to the path metric Λ(S n-1 ) in the state S n-1 at time n-1 to obtain the path metric Λ(S n ¦S n-1 ) of each candidate sequence. Such path metrics are calculated for M states out of the M Q states S n-1 at time n-1 and compared in terms of magnitude to select one path which is most likely to reach one of the states S n at time n. Let it be assumed that, as shown in Fig. 2, known Q-symbol sequences are periodically inserted every N data symbols to avoid uncertainty of the phase of the decoded data. Even if Q=1 is set as the known sequence, no significant performance degradation occurs. The decoding will be described below in detail. (a) To select paths most likely to reach the states S n at time n from the M Q states S n-1 at time n-1, received signal sample sequences z n-q ; q=0, 1, 2, ..., Q-1, Q, are inversely modulated, in accordance with Eq. (6), by phase sequences  n-q ; q=0, 1, 2, ..., Q-1, Q which constitute the states S n and S n-1 , then the inner product of the inversely modulated results with q=0 and the results of summation from q=1 to Q is calculated, and its real part is calculated to obtain a branch metric λ n (S n-1 →S n ) that indicates the likelihood of the transition from the state S n-1 at time n-1 to the state S n at time n. In other words, the multiplication of z n-q by exp(-j n-q ) is equivalent to the inverse modulation of z n-q by  n-q ; the phase sequences  n-q (q=1, 2, ..., Q) constituting the states S n-1 are used to inversely modulate the corresponding ones of the received signal sample sequences z n-q , q=1,2, ....,Q, and the value η n obtained by adding together the inversely modulated results is the results of calculation by Eq. (6). The value η n is used as a reference signal for the sample z n at time n. By multiplying the received sample z n by exp(-j n ) to inversely modulate z n by  n , then the inner product of the resulting value and the value η n of Eq. (6) is calculated, and its real part is calculated to obtain the branch metric by Eq. (5). (b) The branch metric λ n (S n-1 →S n ) is added to the branch metric Λ(S n-1 ) in the state S n-1 at time n-1 to obtain the branch metric of each candidate sequence which passes through the state S n-1 . (c) The above calculations are repeated for all of the M states S n-1 which reach one of the states S n at time n, by which path metrics for M candidate sequences are calculated, and the path metrics are compared in terms of magnitude to obtain the state S n-1 ' which provides the maximum path metric. This state is stored in a path memory as the state of a path at time n-1 which is most likely to reach the states S n at time n, and at the same time, its path metric Λ(S n ¦S n-1 ') is stored as a path metric Λ(S n ) in the states S n at time n. (d) The above calculations are repeated for all the M Q states at time n to obtain survival paths and their path metrics. In this instance, however, since the state of the known Q-symbol sequence is specified as the only one state composed of Q symbols, only a path that reaches the above-said only one state is regarded as a survival at the time of receiving the known pilot sequence (consequently, only one path survives at this time) and the others are excluded. (e) The path metrics in all of the M Q states at time n are compared in terms of magnitude to obtain a state S n '. then the path memory is traced back for a fixed number D of points in time from the state S n ', and the phase  n-D that constitutes the state S n-D reached is outputted as a decoded symbol  n- D , where D is an integer equal to or smaller than N. FIRST EMBODIMENT Fig. 3 illustrates an example of a receiver embodying the method of a first embodiment of the invention. A received wave r(t) from an input terminal 11 is fed to a quasi-coherent detector 12, wherein it is quasi-coherently detected by being multiplied by a local signal from a local signal oscillator 13, and the resulting received signal z(t) of the intermediate frequency (or base band) is sampled by a sampling circuit 14 with a fixed period (a symbol period T) to obtain a complex sample Z n of the received signal z(t). The complex sample is provided to a reference signal generating part 15, wherein it is inputted into a series circuit of delay circuits 15D1 to 15DQ each having a delay T, from which 1- to Q-symbol delayed samples {z n-q ; q=1. 2, ..., Q} are provided to an inverse modulating part 15A. The inverse modulating part 15A calculates the reference signal η n and applies the reference signal to a branch metric calculating part 16. In this embodiment, the states at each time n are defined, for example, as M 2 states that are represented by two phases ( n ,  n-1 ) at time n and n-1 immediately preceding it. When M=4, the trellis diagram of Fig. 1, which represents the state transition in this embodiment, is equivalent to the case where M 2 =16 in Fig. 1, showing that survival paths are decided up to time n-1 and how survival paths to respective states at time n are decided. The branch metric calculating part 16 calculates the branch metric that indicates the likelihood of transition from the state S n-1 at the immediately preceding time to the state S n at time n. The Viterbi decoding part 17 sequentially estimates transmitted phase sequences by the Viterbi algorithm. The reference signal adaptive prediction part 15 predicts the reference signal from which variations in the received signal by fading has been removed. The first embodiment is characterized by the estimation of the transmitted phase sequence by the Viterbi algorithm using, as the branch metric, a square error between a reference signal η n and the received signal sample z n phase-rotated by - n and by the adaptive prediction of the reference signal from the received signal sample sequence. The Viterbi decoding part 17 uses the branch metric to calculate, for each state at time n, the path metric that indicates the likelihood of the sequence which reaches the state, then selects the state at the immediately preceding time from which the path most likely to reach each state at time n originates, and stores, for each state, the path history and the path metric in the path memory 17p and the metric memory 17m, respectively. And, the Viterbi decoding part traces back the path of the minimum one of the path metrics in the M 2 states at time n by the fixed number D of point in time and outputs the decoded symbol phased  n-D . The sequence estimation algorithm according to the present invention described above involves such steps as listed below. (a) At time n there are M 2 states S n represented by two consecutive phases ( n ,  n-1 ). When the Viterbi decoding part 17 selects from the sates S n-1 at time n-1 the state transition that is most likely to reach one of the states S n at time n, the reference signal adaptive prediction part 15 represents the reference signal by the following equation using a real-number prediction coefficient α and calculates it by linear prediction. η n =(1+α)z n-1 exp(-j n-1 )-αz n-2 exp(-j n-2 ) (b) Next, the branch metric calculating part 16 uses a signal phase-rotated Φ n from the received signal sample z n and calculates the following square error between the phase-rotated received signal sample z n and the reference signal η n as the branch metric λ(S n-1 →S n ) indicating the likelihood of the transition from the state S n-1 at time n-1 to the sate S n at time n. λ(S n-1 →S n )=|z n exp(-j n )-η n | 2 (c) The branch metric λ(S n-1 →S n ) is added to the path metric H(S n-1 ) in the state S n-1 at time n-1 to obtain the path metric H(S n ¦S n-1 ) of a candidate sequence that passes through the state S n-1 . (d) The above calculations are repeated for all of the M 2 states S n-1 at time n-1 to obtain the path metrics for M 2 candidate sequences, which are compared in terms of magnitude to obtain the state S n-1 ' of the minimum path metric. This state is stored in the path memory 17p as the state at time n-1 of the survival path which reaches the state S n at time n, while at the same time its path metric H(S n ¦S n-1 ') is stored in the metric memory 17m as the path metric H(S n ) in the state S n at time n. (e) The above calculations are repeated for all of the M 2 states at time n to obtain M 2 path metrics, which are compared in terms of magnitude to obtain the state S n ' of the minimum path metric. The path memory 17p is traced the fixed number D of points in time back from the state S n ' and the phase  n-D that constitutes the state thus reached is outputted as the decoded symbol phase  n-D . SECOND EMBODIMENT In the above-described first embodiment there are M 2 survival paths at each time, but in the second embodiment each state S n at time n is represented by one phase  n . By this, M survival paths are provided at each point in time. The basic decoding operation by the Viterbi algorithm is the same as in the Fig. 3 embodiment, but the reference signal η n is calculated by the following equation using the phase at the immediately preceding time on the survival path whose final state is  n-1 . The receiver configuration in this instance is shown in Fig. 4, in which instead of generating all candidates for the two phases ( n-1 ,  n-2 ) representing the M 2 states at time n-1 in the Fig. 3 embodiment, the phase  n-1 and the subsequent phase at time n-2 on the survival path are read out of the path memory 17p of the Viterbi decoding part 17 and inputted into the inverse modulating part 15A. The inverse modulating part 15A uses the phases  n-1 and n-2 and the received signal samples z n-q and z n-q to calculate the reference signal η n by Eq. (9). The other operations are the same as in the Fig. 3 embodiment. In the first and second embodiments, the prediction coefficient α for predicting the reference signal for use at time n can be adaptively determined in such a manner as to minimize the error between the received signal sample and the corresponding linearly predicted value, for example, by tracing back the phase sequence on the survival path which is connected to each state S n-1 at time n-1. As the result of this, each survival path has its own prediction coefficient. Letting the sequence on the survival path be represented by  n-1 -i (where i=0, 1, ..., n-1), the prediction coefficient α( n-1 ) for predicting the reference signal that is used at time n is selected in such a manner as to minimize a mean square error of an exponential weight that is given by the following equation: where β is a forgetting factor equal to or smaller than 1. η' n-1-i is a reference signal at time n-1-i predicted using the same prediction coefficient α( n-1 ) at all previous points in time and it is given by the following equation: The prediction coefficient α( n-1 ) that minimizes Eq. (11) is as follows: This can be obtained sequentially. That is, where Ω 0 ( 0 )=δ is a small positive number and Θ 0 ( 0 )=0 and z- 1 =0. In Fig. 5 there are indicated by the curves 6C and 7C the results of computer simulations of the error rate performances under no fading environment in the cases where the prediction coefficient α is determined by Eq. (10) with β=1 and the first and second embodiments are applied to the four-phase PSK. The abscissa represents the signal energy per bit versus noise power density ratio E b /N o . For comparison, there are also shown ideal coherent detection performance CD and the differential detection performance DD. The performance 6C by the first embodiment is close to the ideal coherent detection performance within about 1 dB. The difference between the performance 7C by the second embodiment and the ideal coherent detection performance is around 1.5 dB. The amount of processing involved in the second embodiment is approximately 1/4 that in the first embodiment. In Fig. 6 there are indicated by the curves 6C and 7C the performance of the first and second embodiments under the Rayleigh fading environment. f D T represents the speed of fading variation, f D the maximum Doppler frequency (i.e. moving speed of mobile terminal/wavelength of radio carrier) and T one symbol length. Accordingly, 1/T represents the transmission rate. in the differential detection, even if the mean E b /N o is set large, the error rate approaches a fixed value and would not go down below it. In the first and second embodiments, however, the error rate can be reduced by setting the mean E b /N o large. As described above, in the first and second embodiments the prediction coefficient α can be changed in accordance with the fading environment of the received signal, so that the error rate performance is improved as compared with that in the differential detection, regardless of the presence or absence of the fading phenomenon; hence, the performance can be brought close to the ideal coherent detection performance.

Claims: A maximum likelihood decoding coherent detection method for decoding each N successive samples of an M-phase modulated signal with Q samples of known signal inserted therein periodically, the method comprising the steps of: (a) sampling, at the symbol rate T, a received signal to obtain respective received signal samples z n at a series of time points n; (b) inversely modulating samples of the received signal with respective candidates of phases and calculating a branch metric λ(S n-1 →S n ) indicating the likelihood of transition from a state S n-1 to a state S n by using the inversely modulated samples of the received signal; (c) adding said branch metric λ(S n-1 →S n ) to a path metric H(S n-1 ) in said state S n-1 to calculate a path metric H(S n |S n-1 ) of a candidate sequence which passes through said state S n-1 ; (d) repeating the calculation of said steps (b) and (c) for all states S n-1 of paths which enter said state S n , comparing the thus obtained path metrics in terms of magnitude, and selecting the state S n-1 ' of the minimum path metric most likely to reach a state S n ; (e) storing said state S n-1 ' in a path memory as the state of the path which is most likely to reach said state S n , and storing its path metric H(S n |S n-1 ') in a metric memory as the path metric H(S n ) in said state S n ; (f) repeating the above operation of step (e) for all states S n to obtain path metrics; (g) comparing the path metrics in said states S n in terms of magnitude to obtain a state S n ' of the minimum path metric; and (h) tracing said path memory back from said state S n ', and outputting, as the phases of decoded symbols, the phases along the path; characterized in that Q=2, and: step (b) comprises, inversely modulating the received signal samples z n-q by candidate phases  n-q for q=1 and 2, calculating a reference signal η n by the following equation using a prediction coefficient α as a real number, η n =(1+α)z n-1 exp(-j n-1 )-αz n-2 exp(-j n-2 ), and calculating a square error between said reference signal η n and said received signal sample z n phase-rotated by - n as a branch metric λ(S n-1 →S n ) indicating the likelihood of transition from said state S n-1 to S n ; step (d) comprises, repeating the above calculations of said steps (b) and (c) for all of said M 2 states S n-1 , comparing the resulting path metrics in terms of magnitude to obtain a state S n-1 ' of the minimum value, storing it in said path memory as the state through which a survival path reaches said state S n , and storing its path metric H(S n |S n-1 ') in said path metric memory as the path metric H(S n ) in the state S n ; step (f) comprises, repeating the above operation of said step (e) for all of said M 2 states to obtain path metrics; and step (h) comprises, at each time point n, tracing said path memory back for a fixed number D of points in time from said state S n ', and outputting, as a phase of decoded symbol, , the phase  n-D which constitutes part of said state S n-k reached. The method of claim 1, wherein said state S n is represented by only one phase  n , and in said step (b), said reference signal η n is calculated by the following equation: using said prediction coefficient α and the phase , on a survival path which ends with one of said states S n-1 . The method of claim 1 or 2, which further comprises the step wherein, when a reference signal for obtaining a survival sequence to each state S n is assumed, said prediction coefficient α, which minimizes an error between said received signal sample and said reference signal, is calculated by tracing back a decoded sequence by a sequential error minimizing algorithm. The method of claim 1 or 2, wherein said reference signal calculating step is performed by replacing said received signal sample z n-2 with said reference signal η n-1 .

Inventors: ADACHI FUMIYUKI, ADACHI, FUMIYUKI, Adachi, Fumiyuki c/o NTT DoCoMo inc.Pat,&amp; Tradem.
Applicants: NIPPON TELEGRAPH & TELEPHONE, NTT MOBILE COMMUNICATIONS NETWORK INC.
Application year: 1995
Main classification: H04L27
IPC codes: H04L25, H04L27, H04L1