Phase modulation having individual placed edges

Abstract: A phase modulator circuit and method for generating an output signal having individually positionable edges is described. The phase modulator includes a programmable pulse generator, such as an interval counter, or a delay for producing the output signal, and a control value source, such as a memory, for delivering a sequence of control values to the generator. The control values determine the time between successive output pulses.

Description: Background of the Invention Clock signals are square waves used for timing digital systems. Spurious phase modulation of clock signals can lead to problems in system operation. Therefore, it is useful to simulate this spurious modulation with a clock phase modulator to test system performance under stressed conditions. One technique for phase modulation is illustrated in Figure 1. A phase-locked loop (PLL) 10 includes a voltage-controlled oscillator (VCO) 12 that is locked to a carrier signal v c from an oscillator 14. A modulating signal v m added after a phase detector 16 causes the output signal v o to be phase-modulated. One disadvantage of the PLL method is the limited dynamic range of the phase detector 16 and its possible non-linearity. This limits the amplitude of the phase modulation and may cause amplitude distortion. Another disadvantage is the limited frequency range of the closed-loop design and its likely non-flat frequency response. This limits the frequency of the phase modulation and possibly causes frequency distortion. A more thorough treatment of phase modulation using PLLs is discussed in Phase-Locked Loop Circuit Design, by D. H. Wolaver, Section 9-1 (Prentice-Hall, 1991). Another known technique for phase modulation is a direct digital synthesizer (DDS) as shown in Figure 2. A DDS 18 may have a numerical input N m that controls the phase of the output signal v o . At each pulse of a clock signal v c from an oscillator 19, a number N f is added to an accumulator 20, forming a number N p representing the phase. A number N m proportional to the phase modulation is added to N p at summer 22. The sum N p + N m serves as the address for a look-up table 24 that provides a number N s . N s is the value of a sine wave for the phase represented by the sum N p + N m . A digital-to-analog converter (DAC) 26 and a low-pass filter (LPF) 28 produce a sinusoidal voltage V o proportional to N s . The Stanford Telecommunications STEL-2173 device is an example of such a DDS. A disadvantage of the DDS method is the complexity of the circuitry and its resulting high cost, especially for high-speed operation. In particular, the accumulator 20, the sine wave look-up table 24, and the DAC 26 each require large circuits. The size of the circuitry results in delays that slow the speed of operation. Therefore, high-speed operation requires expensive technology such as gallium arsenide semiconductors. US 4,078,192 relates to a synthesizer for multiphase waveforms having a plurality of bistable elements for control of a two-phase induction motor. A clock pulse counter is used to time the setting and resetting of the bistable elements. EP-A-0 021 296 relates to a tone generator for generating signal frequencies in a multiple frequency tone dialling method. The tone generator comprises a pulse oscillator, a frequency divider and counters. Summary of the Invention The invention provides a phase modulator as set out in claim 1. The invention also provides a method of phase modulating a waveform as set out in claim 5. Brief Description of the Drawings The foregoing and other objects, features, and advantages of the invention will be apparent from the following more particular description of preferred embodiments of the drawings in which like reference characters refer to the same parts throughout the different views. The drawings are not necessarily to scale, emphasis instead being placed upon illustrating the principles of the invention. Figure 1 is a block diagram of a prior art phase modulator with a phase-locked loop. Figure 2 is a block diagram of a prior art phase modulator using a direct digital synthesizer. Figure 3 is a block diagram of an example of a phase modulator. Figure 4 is a timing diagram illustrating the edge positions in time produced by a system. Figure 5 is a block diagram of an embodiment of the present invention. Figure 6 is a block diagram of another example of a phase modulator. Figure 7 is a timing diagram of the modulated output signal produced by the phase modulator of Figure 6. Figure 8 is a block diagram of the example of Figure 6 with a flip-flop to position the rising and falling edges. Figure 9 is a timing diagram of the modulated output signal produced by the phase modulator of Figure 8. Figure 10 is a block diagram of a preferred embodiment of the present invention. Figure 11 is a timing diagram of the fine resolution achievable with the phase modulator of Figure 10. Figure 12 is a block diagram of an example without a count-down circuit. Figure 13 is a timing diagram of the modulated output signal produced by the phase modulator of Figure 12. Figure 14 is a schematic block diagram of a programmable delay comprising two delay lines. Figure 15 is a block diagram of the programmable delay circuit of Figure 12. Figure 16 is a timing diagram of the modulated output signal produced by the programmable delay circuit of Figure 15 having correct transition timing and bounded phase slope. Figure 17 is a timing diagram similar to Figure 16 except the phase slope is unbounded. Figure 18 is a timing diagram similar to Figure 16 except the transition timing is incorrect. Figure 19 is a timing diagram of the modulated output signal produced by the programmable interval counter of Figure 6 having correct transition timing. Figure 20 is a timing diagram similar to Figure 19 except the transition timing is incorrect. Detailed Description of the Invention The preferred system of embodiments of the present invention modulate a square wave by governing the position of each edge of the output signal rather than governing the phase or frequency directly. The result is the same as phase or frequency modulation, but the method is simpler and, therefore, faster and less expensive. It uses edge-control circuits in an architecture that allows individual placement of each edge based upon a sequence of control values provided sequentially in response to the output signal. The control values may correspond to the time difference between successive edges of the output signal. They may be, or may be obtained from, values representing the time difference between an edge of the output signal and a corresponding edge of the input signal. Figure 3 illustrates a phase modulator 100 which has a programmable pulse generator 110 and a control value source 112. An external oscillator 114 generates an input signal v c with frequency f c which is an input to programmable pulse generator 110. Programmable pulse generator 110 generates an output waveform v o . Control value source 112 outputs a control value signal 116 to programmable pulse generator 110. The programmable pulse generator 110 uses the input signal v c and the control value signal 116 from the control value source 112 in order to generate the output waveform v o . Output waveform v o can be generated so that its edges (rising, falling, or both) are individually positioned relative to each other or to the input signal v c as determined by the value of the control value signal 116. The control value source 112 outputs a new control value on the control value signal 116 in response to an edge of the output signal v o which forms an input to the control value source 112. The control values output by the control value source 112 may form an arbitrary or ordered sequence such that the result is a square wave clock signal v o with edges placed in any desired pattern. Since the transition to a new control value is made in response to the edges of v o , the generation of spurious, misplaced, or missing edges in the output v o may be avoided as described further below. The sequence of control values is dynamic because the rate at which control value transitions occur is on the order of the nominal frequency of edges in the output signal; preferably, control value transitions occur with each edge. Since the sequence of control values is dynamic, the phase modulation achieves wide bandwidth and wide dynamic range. The circuitry is also simpler and cheaper than that of the prior art. For a given phase modulation, it is possible to calculate the positions in time of the edges of a square wave as illustrated in Figure 4. An algorithm for this calculation is given below. Let the position of the n th edge be t n , which is a sequence of monotonically increasing times. To bound the control values, it is usually convenient to calculate the difference Δ t n between each t n and its position if the square wave had not been phase-modulated. Let the constant interval between unmodulated edges be 1/ f c . Then Δ t n = t n - n / f c If time is quantized into increments of T, these edge positions can be stored in a memory as values N n , where N n = ▵t n / T rounded to the nearest integer. A new value may be provided from memory for each edge. Where the programmable pulse generator 110 is a delay circuit, the value N n can be read from memory and used directly to control the delay of an input pulse and thus locate an output edge. However, in accordance with the present invention the pulse generator is an interval counter which counts input pulses to locate an edge some number of input pulses after a prior edge. In that case, the values N n may be converted into values M n representing the difference between succeeding N n : M n = N n - N n-1 . The M n values may then be added to a constant K, where K = f c /f o , resulting in values L n representing the spacing t n -t n-1 between successive edges of the output signal: L n = ( t n - t n -1 )/ T = (Δ t n - Δt n -1 +1/ f c )/ T = N n - N n -1 + K L n can be read from a control value memory and applied to a programmable interval counter to control edge positions in time. Alternatively, the N n values can be read from memory, and the values can be converted into edge positions in time ( L n ) using simple circuitry that is fast and inexpensive. In the embodiment of the invention of Figure 5, the programmable pulse generator comprises both a programmable interval counter 210 and a programmable delay 220. The control value source comprises memory devices 214 and 224 and may include numerical processors 212 and 222. The Programmable Interval Counter generates pulses spaced L cycles of the oscillator apart. A Programmable Delay changes its delay in response to N1. The advantage of using a programmable interval counter is that it allows a wide range of a nominal frequencies in v o . The programmable interval counter divides down the input frequency f c of input signal v c , so the output frequency can be anywhere from f c divided by the maximum interval of the counter up to f c itself. However, to provide high phase resolution or a high output frequency, a very high speed clock would be required. The use of a programmable delay allows for a higher phase resolution. On the other hand, if only the programmable delay line were used, the length of the delay line might have to be too great to provide a sufficient dynamic range. The delay may even provide a frequency output higher than that of the input clock by sequentially decreasing the delay time as an input clock pulse passes through the delay. This results in multiple output pulses from the delay for each input pulse. In the discussion of frequency and phase, it should be recognized that any change in frequency can be seen as a change in phase and any change in phase can be seen as a change in frequency. Accordingly, a phase modulator can be seen to modulate both frequency and phase. The memories of Figure 5 are stepped in response to the delayed outputs they respectively produce. The numerical processing may be omitted. The memories 214, 224 storing the N1' n and the N2 n may share the same physical device and even share the same addresses. It is usually preferable to have Memory 224 step to the next address in response to the signal v 1 , but it may be simpler in some cases to have it respond to signal v o instead. Similarly, it is usually preferable to have Memory 214 step to the next address in response to the signal v o , but it may be simpler in some cases to have it respond to signal v 1 instead. The result is a square-wave clock signal v o with edges placed in any desired pattern. In Figure 6, an input signal v c from an oscillator 32 has a frequency f c chosen so that f c = 1/T, and f o = f c /K , where f o is the nominal frequency of the output signal, v o . The values N n located in memory 38 are converted by differentiator 40 into values M n representing the difference between succeeding N n : M n = N n - N n-1 . The values M n are added to a constant K (where K = f c / f o ) by summer 42, resulting in values L n representing the spacing t n - t n-1 between successive edges of the output signal as shown in the equations above. Each L n is loaded into a count-down circuit 34, such as a Motorola MC10136 programmable interval counter, which is clocked at a frequency of f c = 1/T. After L n clock cycles, the count-down circuit 34 reaches the terminal count (one), the output of which steps an address counter 36 on counter signal 35 for fetching the next N n in memory 38. The output signal v o also prepares the count-down circuit 34 for a load on the next clock cycle on load signal 37. It should be noted that there are known techniques for phase shifting that use count-down circuits. In one technique, the output signal is used to prepare a count-down circuit for loading a delay value L . This loading alone, however, does not change the value of L. Any changes in the delay value L are made independent of the output signal, unlike the present invention which steps a sequence of control values in response to the output signal. Thus, the known techniques produce relatively static phase shifting and do not achieve the high frequency, dynamic phase modulation of the present invention. The modulated square wave output v o consists of the terminal counts output from the count-down circuit 34, as shown in Figure 7. The terminal counts, which are spaced T x L n apart, are represented here as narrow pulses. These pulses can be widened easily by a one-shot multivibrator or by another count-down circuit (not shown). The implementation in Figure 6 positions only the rising edges. Both rising and falling edges can be positioned by using v o to clock a toggle flip-flop as shown in Figure 8. The alternate example of Figure 8 operates in the same manner as the example of Figure 6, except that the output of count-down circuit 34 serves as the input to toggle flip-flop 39. As shown in Figure 9, the rising and falling edges of the flip-flop output signal v o ' are positioned based upon the rising edges of signal v o . The resolution of the phase modulator 30 shown in Figure 6 is T = 1/f c . For the resolution to be less than a nanosecond, the oscillator frequency must be greater than 1 GHz, which requires more expensive circuitry. A less expensive way to make the resolution smaller without increasing f c is shown in Figure 10. In this preferred embodiment of the invention, a phase modulator 50 achieves fine resolution by using a programmable delay 52 with delay τ n = T1 x N1 n , where the fine-resolution values N1 n are preferably stored in a memory 60. The coarse resolution is achieved with a count-down circuit 54 clocked at f c = 1/T2, based on the coarse-resolution values N2 n also stored in the memory 60. These values are derived from the original position values N n by: N n = Δt n / T1 N1 n = N n mod R , N2 n = ( N n - N1 n ) / R . where R=T2/T1 , and the "mod" operation adds or subtracts R from N n until N1 n lies in the range from 0 to R - 1. Then N n = R x N2 n + N1 n . Since the stored values are binary, this calculation is made easier if R is a power of two. Here, an input signal v c from an oscillator 62 has a frequency f c chosen so that f c = 1/T2. The values N2 n located in memory 60 are converted by differentiator 56 into values M n representing the difference between succeeding N2 n : M n = N2 n - N2 n-1 . The values M n are added to a constant K by summer 55, where K = f c /f o , resulting in values L n representing the spacing between successive edges of an intermediate signal v 1 . Each L n is loaded into the count-down circuit 54, which is clocked at a frequency of f c = 1/T2. After L n clock cycles, the count-down circuit 54 reaches the terminal count, which steps an address counter 58 on counter signal 57 and prepares the count-down circuit 54 for a load on the next clock cycle on load signal 59. The address counter 58 steps the memory 60 to output the next values N2 n and N1 n . The modulated intermediate signal v 1 consists of the terminal counts output from the count-down circuit 54 and are spaced T2 x L n apart, as shown in Figure 11. Referring again to Figure 10, the intermediate signal v 1 forms an input to the programmable delay circuit 52. The falling edge of each delayed terminal count on register signal 63 steps the next N1 n from register 64 for producing the delay T n = T1 x N1 n in the programmable delay circuit 52. The fine resolution T1 can be made arbitrarily small, limited only by the accuracy and stability of the delays. In a variation of this embodiment, the difference and summation processes can be performed before the values are stored in memory; the L n can be stored rather than the N2 n . An algorithm to determine edge positions will now be described. For an unmodulated square wave, the phase p(t) in cycles increases linearly with time: p(t) = f o t. The phase of a modulated square wave is p(t) = f o t +Δp(t), where Δ p(t) is the desired phase modulation in cycles. A rising edge will occur when p(t) is an integer (0, 1, 2, ...). Then the time t n of the nth rising edge is found by solving Eq. 3 for t n . f o t n + Δp(t n ) = n , Δt n = t n - n / f o . This equation cannot be solved in general, and the t n must be found by trial and error. However, a good approximation to t n can be found with an iterative equation. Without loss of generality, let p(0) = 0. Then Δf ( t ) ≡ d dt Δp ( t ), t 0 =0, t n +1 = t n + 1 f o + Δ f ( t n + 0.5/ f o ) , Δ t n = t n - n / f o . Taking a specific example of nominal frequency f o = 16 and a sinusoidal phase modulation of Δ p(t) = (6/2π)·sin(2 πt ), then Δ f(t) = 6·cos(2 πt ), and the Δ t n found from Eq. 4 are shown in the table below. Also shown in the table are the ideal values for Δ t n determined from Eq. 3. For this case, the difference between the Δ t n determined from Eq. 4 and the ideal Δ t n is less than 6%. If the resolution is T = 0.0001, then the numbers stored in memory are found by Eq. 1 (see the N n in the table). If the phase modulator is to be implemented with a delay line as in Figure 10 with T1 = 0.0001, T2 = 0.0005, R = 5, and K = 125, then the fine-resolution numbers and the coarse-resolution numbers are found by Eq. 2 (see N1 n and N2 n in the table). n Δt n (ideal) Δ t n N n N1 n N2 n 0 0.0000 0.0000 0 0 0 1 -0.0169 -0.0168 -168 2 -34 2 -0.0327 -0.0324 -324 1 -65 3 -0.0463 -0.0456 -456 4 -92 4 -0.0560 -0.0548 -548 2 -110 5 -0.0597 -0.0581 -581 4 -117 6 -0.0538 -0.0524 -524 1 -105 7 -0.0340 -0.0340 -340 0 -68 8 0.0000 -0.0012 -12 3 -3 9 0.0340 0.0352 352 2 70 10 0.0538 0.0570 570 0 114 11 0.0597 0.0629 629 4 125 12 0.0560 0.0586 586 1 117 13 0.0463 0.0481 481 1 96 14 0.0327 0.0340 340 0 68 15 0.0169 0.0178 178 3 35 16 0.0000 0.0008 8 3 1 If either |Δ p(t) |<<1 or |Δ f |<< f o , then there is an even simpler algorithm to approximate Δ t n : Δ t n = -Δ p ( n / f o ) f o If the phase slope is bounded by | N2 n - N2 n-1 |< K /2, then N n can be reduced to conserve memory. Replace N2 n with N3 n = N2 n mod K , and replace the operation L n = N2 n - N2 n-1 + K with: L n = N3 n - N3 n -1 +2 K for N3 n - N3 n -1 <- K /2 L n = N3 n - N3 n -1 + K for -K /2≤ N3 n - N3 n -1 < K /2 L n = N3 n - N3 n -1 for K /2≤ N3 n - N3 n -1 . This operation is easily implemented when K = 2 k by performing the subtraction in k-bit signed numbers and always adding only K. When the desired output frequency f o is very high, it is desirable to make the oscillator frequency f c equal f o to avoid an expensive high-frequency count-down circuit. Then all of the phase modulation is done with the programmable delay, and N n = N1 n (there is no N2 n ). Let T2 ≡ 1/ f o , and R ≡ T2/T1 , where T1 is the resolution of the delay. If the phase slope is bounded by | N1 n - N1 n-1 |<R /4, then N1 n can be reduced to conserve memory and to reduce the size of the programmable delay. Replace N1 n with N3 n = N1 n mod R. An example without a count-down circuit is shown in Figure 12. It is usually convenient for the duty cycle of v 1 to be 50%, especially for higher frequency operation to allow for greater range. The falling edge of the delayed signal v o output from a programmable delay 74 advances an address counter 76 after an additional delay of T2 /4. Given the bound on the phase slope, this additional delay assures that the waveform of v o is low both immediately before and immediately after the change in N3 n . The address counter 76 steps the memory 78 to output the next N3 n . The waveforms for v 1 and v o shown in Figure 13 are for an example where R = 8 and the N3 sequence is 2, 1, 0, 7, 6, 5, 4. Note that the third rising edge of v 1 appears twice in v o , delayed first by T1N3 2 = 0 and then by T1N3 3 = T1 x 7. If the programmable delay 74 has significant set-up time (compared to T2 ) before the delayed waveform is available after applying a new N3 n , then two programmable delays may be necessary. Figure 14 illustrates a possible implementation of alternating delay line circuit 75 in place of block 75 from Figure 12. The delay lines 74a and 74b each receive the input signal v 1 and the control values N3 n . The delay line 74a handles the odd-ordered N3 n while the delay line 74b handles the even-ordered N3 n . The next succeeding control value N3 n is placed in registers 82a or 82b for being alternately loaded into the respective delay lines 74a and 74b. Output logic comprising AND gates 84a and 84b, OR gate 86, and type D flip-flop 88 performs a multiplexing function on the output of the delay lines 74a and 74b. The output logic also controls the alternating operation of the delay lines. Specifically, the delay line output signal 83a forms an input to AND gate 84a and delay line output signal 83b is an input to AND gate 84b. The Q and Q output signals of flip-flop 88 form the other inputs to AND gates 84a and 84b. The OR gate 86 combines the output of AND gates 84a and 84b to form the modulated output signal v o . The output of OR gate 86 is also inverted to serve as the clock input to flip-flop 88. With each cycle of input signal v 1 , the delayed outputs of delay lines 74a and 74b are alternately selected through the operation of flip-flop 88. The outputs of AND gates 84a and 84b are also inverted to enable respective registers 82a and 82b. As noted above, the aspect of the invention whereby the transition to a new control value is made in response to the edges of the output signal allows the generation of spurious, misplaced, or missing edges in the output signal to be avoided. Having described several embodiments of the present invention, the following discussion focuses on the additional conditions required to avoid the undesired edges in the output signal. Figure 15 shows the detail of the programmable delay 74 of Figure 12. The programmable delay 74 includes a series of delay sections, D, each delay section adding a fixed delay of T2/16, where T2 = 1/ f o , to the input signal v 1 . A series of taps, P0 through P15, are coupled in parallel between the delay sections D and a multiplexer 73 to produce the output signal v o , dependent on the delay value N3 n input to the multiplexer 73. The timing diagram of Figure 16 shows the relationship between the delay taps, P0 to P15, and the output signal v o of the delay 74. As can be seen from the diagram, the value N3 n changes a time T2/4 after the falling edge of v o . Since the phase slope is bounded by | N3 n - N3 n-1 | <R/4, where R = T2/T1 and T1 is the resolution of the programmable delay, the rising edges of v o are correctly placed. In contrast, Figure 17 illustrates the undesirable result produced when the phase slope is not bounded. An unbounded phase slope, i.e., too large a phase change, results in both spurious and misplaced edges in the output signal v o . A similar undesirable output results when the transition time to a new control value is incorrect. The correct transition time assures that the output signal v o is low both immediately before and immediately after the change in N3 n . Figure 18 shows the spurious and misplaced edges resulting from incorrect transition times of T2/8 and T2/2.7, for example. An example in which the programmable pulse generator is a count-down circuit was shown earlier in Figure 6. The timing diagram of Figure 19 shows that such an embodiment correctly places edges of the output signal v o when the control values L n are changed a transition time T/2 after the falling edge of v o . The transition time of T/2 allows for greatest flexibility in changing phase. In contrast, Figure 20 illustrates the undesirable results produced when the control values L n are changed independent of the output signal v o . In this example, the values L n are changed instead every 5T, where T is the period of the oscillator signal v c . The timing diagram of Figure 20 shows an extra "spurious" edge produced when L n = 3. An edge associated with L n = 4 is missed because by the time the count-down circuit has finished counting down from the previous value of L n-1 = 8, the next value, L n+1 = 5, has been loaded. The previous examples illustrate that making the transition timing of the control values responsive to the output signal of the programmable pulse generator is necessary to be able to avoid spurious, misplaced, or missing edges. In addition, phase slope bounding and correct transition timing are also necessary to achieve desirable results.

Claims: A phase modulator for modulating a waveform comprising: a programmable interval counter (210,54), coupled to receive the waveform, for producing a pulsed intermediate signal wherein the position of each edge of the intermediate signal relative to the waveform is determined in response to a first control value provided to the programmable interval counter (210,54); a first control value source (214,60) coupled to the programmable interval counter for producing a sequence of first control values wherein the next first control value in the sequence is provided to the programmable interval counter in response to an edge of the intermediate signal; a programmable delay circuit (220,52), coupled to receive the intermediate signal, for producing a pulsed output signal wherein the position of each edge of the output signal relative to the intermediate signal is determined in response to a second control value provided to the programmable delay circuit (220,52); and a second control value source (222,60) coupled to the programmable delay circuit (220,52) for producing a sequence of second control values wherein the next second control value in the sequence is provided to the programmable delay circuit (220,52) in response to an edge of the output signal. The phase modulator of claim 1 wherein the first and second control value sources comprise a common memory (60). The phase modulator of any preceding claim wherein the waveform comprises a square wave signal produced by an oscillator (202,62,72). The phase modulator of claim 1 wherein the first control value source (214,60) includes a numerical processor (212) for converting a first value, corresponding to the time difference between the position of an edge of the intermediate signal and its position if the waveform had not been modulated, to a second value corresponding to the time difference between successive edges of the intermediate signal, the second value being the first control value. A method of phase modulating a waveform comprising the steps of: receiving the waveform; producing a pulsed intermediate signal using a programmable interval counter (210,54) wherein the position of each edge of the intermediate signal relative to the waveform is determined in response to a first control value; producing a sequence of first control values wherein the next first control value in the sequence is provided in response to an edge of the intermediate signal; producing a pulsed output signal using a programmable delay circuit (220,52) wherein the position of each edge of the output signal relative to the intermediate signal is determined in response to a second control value; and producing a sequence of second control values wherein the next second control value in the sequence is provided in response to an edge. of the output signal.

Inventors: CHANG PAUL, MITCHELL J HOWELL JR, WOLAVER DAN H, CHANG, PAUL, MITCHELL, J. HOWELL JR., WOLAVER, DAN H.
Applicants: TEKTRONIX INC, TEKTRONIX, INC.
Application year: 1995
Main classification: H03M1
IPC codes: H03C3, G06F1, H03K7