Decision feedback equalizer for cancelling short- and long-multipath components using correspondingly delayed decision symbols

Abstract: In a decision feedback equalizer, transmitted signals are received by an array of antennas and baseband symbols are recovered. A forward filter provides weighting of the recovered baseband symbols with first coefficients so that precursor multipath components of a transmitted symbol are canceled and combines the weighted symbols. A set of first delay elements provide respective delays to one of the baseband symbols to produce a short- and a long-delayed baseband symbol. A set of correlators is provided for detecting correlations between the baseband symbol and the short- and long-delayed baseband symbols. A set of second delay elements provide respective delays to a decision symbol to produce a short- and a long-delayed decision symbol. A selector is responsive to a correlation detected by one of the correlators for selecting one of the short- and long-delayed decision symbols. A feedback filter shifts the selected decision symbol along a tapped-delay line to produce successively shifted decision symbols, multiplies them with second coefficients and combines the multiplied decision symbols to produce an output signal that cancels postcursor multipath components of the transmitted symbol. The output signals of the forward and feedback filters are combined to produce a pre-decision symbol which is used by a decision circuit to produce a decision output for coupling to the second delay elements as the aforesaid decision symbol.

Description: BACKGROUND OF THE INVENTION Field of the Invention The present invention relates generally to receivers which can be used for digital radio communication systems or sonar detection systems, and more specifically to a decision feedback equalizer with an array of antennas, or sensors for canceling multipath components of a signal received via time-varying multipath-fading channels. Description of the Related Art The combined use of an array of spaced antennas, or sensors with a decision feedback equalizer for canceling multipath-fading related intersymbol interference is known, as shown and described in EP-A 604, 956, published on January 6, 1994. Using the well-known least mean square (LMS) algorithm, the main lobe of the antennas is oriented in the arrival direction of a target signal, while forming a null in the direction of an undesired signal. However, if multipath components are arriving through very long return paths, their delay times are likely to exceed the length of the dapped-delay line of the equalizer, and hence no equalization would result at all. In particular, as the data transmission speed of a channel grows faster, the delay profile of the transmitted signal normalized by modulation speed extends, increasing the severity of the problem. GB 2238932, published on June 12 1991, describes a decision feedback equalizer including a forward part whose signal reference point is shiftable depending on the channel response. A center tap of a transversal filter, which forms part of the forward equalizer, is shifted by at least one symbol interval. An impulse response estimation circuit receives the decision output of the forward equalizer, and this is controlled following the channel response to achieve adaptive equalization. EP 0565104, published October 13 1993, discloses an interference canceller with feedforward transversal filter having switched tap-gain coefficients. An adaptive feedforward equalizer operates on the incoming signals according to a first error signal. A first feedback equalizer then operates on the output of a decision circuit according to the first error signal to cancel postcursor distortion. A second feedback equalizer nonlinearly filters the output of the decision circuit according to a second error signal and generates an estimate of a desired component of the output of the feedforward equalizer. The outputs of the second feedback equalizer and the feedforward equalizer are combined to produce the second error signal, or an estimate of an undesired component of the output of the feedforward equalizer. In the presence of the interference, the desired second error signal estimate is combined with the outputs of the feedforward equalizer and the first feedback equalizer to produce an equalized signal for coupling to the decision circuit. SUMMARY OF THE INVENTION A preferred embodiment of the present invention provides a decision feedback equalizer capable of canceling multipath-fading related intersymbol intereference caused by very long multipath fading channels without increasing the length of the equalizer and canceling interference from jamming sources. A further embodiment of the present invention provides a solution to a collision problem that occurs when multipath components (echoes) of an earlier transmitter symbol arrive at the equalizer simultaneously with those of a latter transmitted symbol. According to a first embodiment of the present invention, there is provided a decision feedback equalizer comprising a plurality of sensors for receiving transmitted signals and recovering baseband symbols from the received signals, a forward filter for weighting the recovered baseband symbols with coefficients so that precursor multipath components of a transmitted symbol are canceled and combining the weighted baseband symbols, a subtractor having a first input for receiving an output signal of said forward filter and a second input for receiving a postcursor canceling signal supplied from a feedback filter means to produce a predecision signal, decision means for producing a decision output from the predecision symbol from the output of said subtractor, and a feedback filter for shifting an input decision symbol along a tapped-delay line to produce successively shifted decision symbols, weighting the successively shifted decision symbols with coefficients and combining the weighted decision symbols to produce said postcursor canceling signal at the second input of said subtractor, characterized by a plurality of delay means for providing delays to a decision symbol to produce a short-delayed decision symbol and long-delayed decision symbol from the output of said decision means, and a correlation detector for detecting the correlation between one of the baseband symbols and said short-delayed and long-delayed baseband symbols, in that said feedback filter is arranged to receive the short-delayed decision symbol, as said input decision symbol, from said delay means if said short-delayed recovered symbol is detected by said correlation detector and receive the long-delayed decision symbol, as said input decision symbol, from said delay means if said long-delayed recovered symbol is detected by the correlation detector. BRIEF DESCRIPTION OF THE DRAWINGS The present invention will be described in further detail with reference to the accompanying drawings, in which: Fig. 1 is a block diagram of a decision feedback equalizer using an adaptive filter array according to an embodiment of the present invention; Fig. 2 is a diagram illustrating a simplified antenna array; Figs. 3A and 3B show antenna patterns obtained respectively by computer simulations of the equalizer of the present invention and a prior art equalizer having no feedback filter; Fig. 4 is a block diagram of a modified decision feedback equalizer of the present invention; and Fig. 5 is a block diagram of a further modification of the present invention. DETAILED DESCRIPTION Referring now to Fig. 1, there is shown a DFE (decision feedback equalizer) receiver according to a first embodiment of the present invention. The receiver is most suitably used for applications such as digital radio communication systems or sonar detection systems where the quality of a transmitted signal is seriously affected by multipath components of the signal. At the transmit site, a symbol sequence {.....a -1 , a 0 , a +1 .....} is quadrature-modulated onto a carrier in a format such as phase shift keying and transmitted. The receiver comprises an array of antennas 10 1 - 10 N spaced at half-wavelength intervals of the transmitted carrier, a plurality of receivers 11 1 - 11 N , and an adaptive array filter 1 formed by a plurality of complex multipliers 12 1 - 12 N and an adder 13. The outputs of antennas 10 1 - 10 N are connected respectively to receivers 11 1 - 11 N where they are quadrature-demodulated, recovering baseband vectors x 1 , x 2 , .... x N . These baseband vectors are applied to complex multipliers 12 1 - 12 N , where they are respectively multiplied by weight coefficients c 1 , c 2 , .... c N supplied from coefficient update circuits 40 1 - 40 N , so that precursors (future symbols) of multipath-fading related intersymbol interference are canceled. The weighted vectors are summed together in combiner 13. The output of the adaptive array filter 1 is applied to a subtractor 14 where it is combined with a postcursor canceling signal supplied from a feedback filter 60 to produce an equalized output vector "y" at the output of subtractor 14. The output of subtractor 14 is coupled to a decision circuit 15 for making a decision in favor of logic-1 or logic-0 depending on the result of a comparison with a threshold value, producing a sequence of estimated symbols {a i }. An error detector 19 is connected across the input and output of decision circuit 15 to detect a decision error ε which is suitably weighted by a correction factor µ in a multiplier 20 and supplied to update circuits 40 1 - 40 N . Each update circuit 40 includes a complex correlator 41 in which the weighted decision error µ.ε is multiplied by the corresponding incoming symbol x i * (where (*) represents the complex conjugate, and i = 1, 2, ....., N). The weight coefficient c i is produced by coupling the output of correlator 31 to the negative input of a subtractor 42 and applying its output through a symbol delay (T) element 43 to the other input of subtractor 42 forming a feedback loop so that it operates according to the LMS (least mean square) algorithm c i n = c i n-1 - µ x i * ε (where n is the time indicator) and hence the mean square value of the decision error ε is reduced to a minimum. Note that the decision circuit 15, decision error detector 19, feedback filter 60 and tap-weight update circuits 50 1 - 50 N form a decision feedback equalizer. Postcursors (previous symbols) of multipath-fading related intersymbol interference are canceled by coupling the output of decision circuit 15 to the feedback filter 60 through a selector circuit 70. The feedback filter 60 comprises an M-stage tapped-delay line having symbol-delay elements 16 1 - 16 M-1 . When a transmitted symbol A 0 is reproduced at the output of decision circuit 15, M past decision output symbols A -1 , A -2 ,.....,A -M appear at the successive taps of the M-stage delay line and are presented to complex multipliers 17 1 , 17 2 ,.....,17 M , respectively, where they are weighted by tap-weight coefficients d 1 , d 2 ,....., d M , respectively. The weighted tap signals are summed in an adder 18 to produce a postcursor canceling signal which is applied to the subtractor 14 to cancel postcursors from the output of adder 13. The tap-weight coefficient d j (where j = 1, 2,.....,M) of feedback filter 60 is produced according to the LMS algorithm d j n = d j n-1 + µA- j * ε . Specifically, the weighted decision error µ.ε is coupled to tap-weight update circuits 50 1 - 50 M . Each update circuit 50 includes a correlator 51 for multiplying the weighted decision error µ.ε by complex conjugate of past decision output symbol A -i , the output of correlator 51 being coupled to one input of an adder 52 whose output is connected through a symbol delay (T) element 53 to the other input of adder 52. The DFE includes a plurality of delay elements 21, 22 and 23. One of the outputs of delay elements 21, 22, 23 is selected by selector circuit 70. Delay elements 21, 22 and 23 are connected to the output of decision circuit 15 to provide a delay time corresponding to T, τ 1 and τ 2 , respectively, to decision symbol A 0. Note that the delay time τ 1 is greater than M ×T, i.e., greater than the total delay time of feedback filter 60. The delay time τ 2 is greater than τ 1 . Delay times T, τ 1 and τ 2 are the delay times of multipath components of a transmitted symbol a 0 and the T-delayed multipath component is a near-field reflection and the τ 1 - and τ 2 - delay times are far-field reflections propagating over long paths. The occurrences of these far-field components are known in advance from experiments. Typically, τ 1 and τ 2 are 100T and 200T, respectively. A plurality of delay elements 27, 28 and 29 are connected to the output of receiver 11 1 to provide delays T, τ 1 and τ 2 , respectively, to the received baseband symbol x 1 . Correlations between the symbol x 1 and the outputs of delay elements 27, 28 and 29 are detected by correlators 30, 31 and 32, respectively. If multipath components of delays T, τ 1 and τ 2 exist for a given transmitted symbol and arrive in sequence, correlators 30, 31 and 32 will sequentially produce their output signals. Selector 70 includes comparators 33, 34 and 35 and gates 24, 25 and 26. Comparators 33, 34 and 35 are connected to the outputs of correlators 30, 31, 32, respectively, for making comparisons with a predetermined reference value, and supply their outputs to gates 24, 25, 26, respectively. If the output of a corresponding correlator exceeds the reference value, each comparator supplies a gate-on pulse to the associated gate for coupling a delayed decision symbol to the feedback filter 60. Assume that a transmitted symbol a 0 has a multipath component arriving at an instant delayed by symbol interval T with respect to the symbol a 0 , a correlation is detected between them by correlator 30 and a gate-on pulse is supplied from comparator 33 to gate 24 to pass the output of delay element 21 to the feedback filter 60. Therefore, a decision output symbol A 0 corresponding to the transmitted symbol a 0 appears at the output of delay element 21 as a past symbol A -1 time-coincident with the detection of the corresponding T-delayed multipath component by correlator 30. This past symbol is applied through gate 24 to the first tap of feedback filter 60, where it is weighted by coefficient d 1 for coupling to adder 18, a producing postcursor canceling output A -1 d 1 at the output of adder 18. Using this canceling signal, the T-delayed multipath component is canceled at the output of subtractor 14. The decision symbol A -1 will then be shifted from delay element 16 1 to the next element 16 2 and appears at the second tap as a past decision symbol A -2 and weighted by coefficient d 2 . If there is a 2T-delayed multipath component associated with the transmitted symbol a 0, the update circuit 50-2 will update the coefficient d 2 using the weighted decision error µ.ε and so the past symbol A -2 is weighted with the updated coefficient d 2 , producing a canceling signal. This canceling signal is time coincident with the 2T-delayed multipath component which appears at the output of combiner 13 at the instant two symbols delayed with respect to the symbol a 0 . A 3T-delayed multipath component, if present, will be canceled in a similar manner by a canceling signal A -3 d 3 . Therefore, multipath components of a given transmitted symbol are canceled in this manner insofar as their longest delay does not exceed the total delay time of feedback filter 60. Quantitative analysis of the cancellation of multipath components whose longest delay does not exceed the total delay time of feedback filter 60 will be given below using simultaneous linear equations. It will also be shown that the adaptive array filter 1 operates in the same manner as a feedforward filter which usually forms part of a decision feedback equalizer. First, the output "y" of subtractor 14 is given in the form: Since the past decision symbol can be approximated as a transmitted symbol if the symbol error rate can be ignored, the decision error ε is given by: ε = y - a 0 The weight coefficients c i and d j that minimize the mean square value of decision output e are determined by a normal equation (Wiener-Hoph Equation) using the orthogonality principle. From the following relations, the normal Equation can be derived, using linear weight coefficients as unknown variables. E[ε·x i *] = 0 (i = 1, 2, ...., N) E[ε·a j *] = 0 (j = -1, -2, ...., -M) where E represents the expectation. A simplified antenna array of the embodiment of Fig. 1 is illustrated in Fig. 2 to derive the normal equation using a simplified dispersive propagation model. Let hi represent the impulse response of a propagation path, with h 0 representing a reference impulse response for the symbol of interest. The impulse responses h -i and h +i represent the precursor and the postcursor, respectively. The received signals are represented by convolution of the transmitted symbol sequence {a i } and the impulse response h i . The main signal component S i received by each antenna is given by h 0 a 0 and a multipath component received by each ancenna is given by h i a 0-i . Because of the 1/2 antenna spacing, the main signal component S i received by antenna 10 i is delayed by an amount equal to exp {-j(i -1) 0 } with respect to the main component S 1 received by the first antenna 10 1 , where  0 is the angle of arrival of the main signal S i to the axis of each antenna and is given by  0 = π sin  0 . Therefore, the baseband vectors x i are represented as follows: where n i is the noise contained in the output of antenna 10 i and  n represents the angle of arrival of a multipath signal component corresponding to the n-th impulse response of the dispersive propagation model and is given by  n = π sin  n . From Equations (3) and (4), the normal Equation is given as follow: where, C and D are in the form ψ pq represents the N x N correlation matrix of signals received by antennas 10 1 - 10 N and the elements of this matrix are given by: where p, q = 1, 2, ....., N, σ 2 is the noise power, δ pq is the Kronecker's delta which equals 1 if π =  or 0 if π ≠ , I is an M x M unit matrix, and 0 is an M-th order zero vector, H is the N x M correlation matrix of the feedback filter 60 given by Equation (9), and S is a correlation vector correlating the symbol vector (x 1 , x 2 , .... x n ) with the decision output symbol as given by Equation (10). Equation (6) indicates that the operation of the antenna array and the feedback filter is represented by simultaneous linear equations and that the antenna array functions as a feedforward filter of the decision feedback equalizer. Therefore, the present embodiment is not simply a sum of, but an integrated combination of, an antenna array and a decision feedback equalizer. A computer simulation was made for evaluating the present embodiment in terms of the directivity pattern of the antenna array which is given by: where, Γ is a unit DC signal vector that is assumed to be arriving at antennas 10 1 - 10 N at an angle q and is given by: Γ T = 1 · [1 exp{-j()}exp{-j2()} .....exp{-j(N-1)()}| where () = π sin . In the computer simulation, a four-element antenna array (each element with a 10 dB signal-to-noise ratio) and a single-delay tap feedback filter were used for a three-wave multipath fading model by representing a main signal as h 0 a 0 , a phase-advanced multipath component as h -1 a +1 , and a phase-lagged multipath component as h +1 a -1 , and setting the amplitude of the impulse responses to a unit value (i.e., |h -1 | = |h 0 | = |h +1 | = 1, which represents the worst situation where the frequency selective fade is dominant) and setting the arrival angle of the main signal ( 0 ), phase-advanced and lagged multipath components ( -1 ) and ( +1 ) as  0 = 20°,  -1 = 45° and  +1 = 20°, respectively. For purposes of comparison, a simulation was further made on a prior art four-element adaptive array receiver having no feedback filter, using the same operating parameters just described. Results of the computer simulations are shown in Figs. 3A and 3B. The antenna pattern shown at Fig. 3A is the result of the simulation derived from a four-element antenna array embodying the present invention. It is seen that the main lobe is oriented in the arrival direction (=20°) of the main signal and a null point (a point of zero gain) is formed in the arrival direction of the phase-advanced multipath component. While the phase-lagged multipath component h +1 a -1 is summed with the main signal, it is cancelled with an estimate of the symbol a -1 derived from the first delay-line tap of the feedback filter. It is therefore seen that, regardless of the presence of multipath components of a desired signal, the directivity pattern of the antenna array is constantly oriented in the arrival direction of the main component of the desired signal so that it is received at a maximum signal-to-noise ratio. On the other hand, the directivity pattern shown at Fig. 3B is the result of simulation derived from the prior art adaptive array receiver. Since the prior art receiver has no feedback filter and provides cancellation by forming null points in the antenna pattern, it is impossible to cancel the phase-lagged multipath component h + 1 a -1 from the main component with which it is received with the same antenna gain. As a result, the reduction of the phase-lagged multipath component would cause the antenna gain to be reduced at the cost of the main signal. If the delay times of multipath components exceed the total delay time of the feedback filter, no equalization proceeds in the feedback filter until correlation is detected again. Under such conditions, the normal equation is given in the form: Such conditions correspond to a receiver having no feedback filter, where an antenna pattern similar to that of Fig. 3B would result. Multipath components whose delay times exceed the total delay time of the feedback filter 60 may be few or nonexistent under normal multipath fading environment. However, very long delay paths may exist in a particular situation such as sonar and the like and multipath components coming through such long paths experience delays equal to or greater than τ 1 . Therefore, if there is a τ 1 -delayed multipath component for a given transmitted symbol a 0 , a correlation is detected between a 0 and the τ 1 component by correlator 31 and a gate-on pulse is supplied from comparator 34 to gate 35 to pass the output of τ 1 -delay element 22 to the feedback filter 60. Therefore, a decision output symbol A 0 corresponding to the transmitted symbol a 0 appears at the output of delay element 22 simultaneously with the time of detection of the corresponding τ 1 -delayed multipath component. This past symbol is applied through gate 25 to the first tap of feedback filter 60 as a past symbol A -τ1 , where it is weighted by coefficient d 1 , producing a postcursor canceling signal A -τ1 ·d 1 at the output of adder 18. Using this signal, subtractor 14 cancels the τ 1 -delayed multipath component. If the τ 1 -delayed multipath component is followed by a series of multipath components at successive T (symbol) intervals, such multipath components will be adaptively canceled as the decision symbol A -τ1 is successively shifted along the tapped-delay line of the feedback filter. This process continues until the decision symbol A -τ1 appears as A -(τ1-M) at the end of the tapped-delay line of the feedback filter. This process can be represented by the following normal equation: where, D' indicates the vector of tap-weight coefficients d i of the feedback filter when the τ 1 -delayed symbol is being shifted along its tapped-delay line, and D indicates the vector of tap-weight coefficients d i of the feedback filter when the T-delayed symbol is being shifted along its tapped-delay line. The vector D no longer exists when D' exists and hence D = 0 in Equation (14). It is seen from Equation (14) that the size of the correlation matrix of Equation (13) is extended from (N + M) x (N + M) to (N + M + τ 1 ) x (N + M+ τ 1 ). Similar events occur when multipath components of a transmitted symbol return at instants delayed by amount equal to τ 2 or greater. The multipath components of an earlier-transmitted symbol may arrive at the same instants as a significant multipath component of the most recent symbol. Under such circumstances, correlations will be simultaneously detected by correlators 30 and 31 to open the gates 24 and 25. When this occurs, equalization cannot proceed in the feedback filter of Fig. 1. A solution to this problem is illustrated in Fig. 4 as a modification of the first embodiment of this invention. According to this embodiment, the modified DFE receiver includes feedback filter 60' and delay elements 22, 23, 28, 29 and correlators 31, 32. Modified selector circuit 70' includes only comparators 34, 35 and gates 25, 26. This embodiment further differs in that the output of decision circuit 15 is further connected through a T-delay element 61 to the first tap of the feedback filter 60' The first and second taps of feedback filter 60' are separated from each other, and the output of selector circuit 70' is connected to the second tap of the filter 60'. Since it is known that a T-delayed multipath component is the dominant component, it is sufficient to take account only of this component in a situation where it is usually likely to collide with a multipath component that propagates over a very long path. The output of delay element 61 is the decision symbol corresponding to such a T-delayed multipath component. The output of this delay element is coupled to multiplier 17 1 where it is weighted by coefficient d 1 , producing a signal for canceling the T-delayed multipath component at the output of subtractor 14. Simultaneously with the arrival of the T-delayed multipath component, a τ 1 -delayed multipath component of a previous symbol is received and correlation is detected by correlator 31, resulting in the gate 25 being open to pass the τ 1 -delayed decision symbol A -τ1 to the second tap of feedback filter 60' where it is multiplied by coefficient d 2 in the multiplier 17 2 , providing a signal for canceling the τ 1 -delayed multipath component at the output of subtractor 14. Multipath components arriving at successive T-intervals following the τ 1 -delayed multipath component are canceled as the τ 1 -delayed decision symbol is shifted along the feedback filter 60'. Similar events occur with respect to a τ 2 -delayed multipath component and those following it. In this case, correlator 32 produces an output which opens the gate 26 to pass the output of delay element 23 to the second tap of filter 60'. In this modified embodiment, the output signal Z of the feedback filter 60' is given as follows: where L corresponds to τ 1 or τ 2 . It is seen that the first term of Equation (15) represents the output of feedback filter 60' for canceling the T-delayed multipath component associated with the transmitted symbol a 0 , the second term representing the filter output when canceling those components associated with the previously transmitted symbol a -τ1 or a -τ2 . If there is a 2T-delayed multipath component of substantial magnitude in addition to the T-delayed component, an embodiment shown in Fig. 5 is suitable. This embodiment is similar to those of the Fig. 4 embodiment with the exception that it includes delay elements 62 and 63 of 2T-delay time, a correlator 64 and a gate 66, Correlator 64 provides detection of a correlation between the symbol x 1 and the output of delay 63 and gate 66 is connected between delay 62 and the second input of feedback filter 60'. Since there is a likelihood of the 2T-delayed multipath component time-coincident with a multipath component associated with an earlier-transmitted symbol a -τ1 or a -τ2 , selector 70" includes a maximum detector 65 for comparing the outputs of correlators 64, 31 and 32 with each other to find which one has the highest correlation value and supply a gate-on pulse to one of gates 66, 25 and 26 corresponding to the correlator that produces the highest correlation value. In this way, the multipath component of highest magnitude is selected in preference over other simultaneously arriving multipath components and the associated delayed decision symbol is selected in preference over other delayed decision symbols by selector circuit 70" and applied to the second tap of feedback filter 60', which cancels the highest magnitude multipath component at the output of subtractor 14.

Claims: A decision feedback equalizer comprising: a plurality of sensors (10 1 -10 N , 11 1 -11 N ) for receiving transmitted signals and recovering baseband symbols from the received signals; a forward filter (1) for weighting the recovered baseband symbols with first coefficients so that precursor multipath components of a transmitted symbol are canceled and combining the weighted baseband symbols; a subtractor (14) having a first input for receiving an output signal of said forward filter (1) and a second input for receiving a postcursor canceling signal supplied from a feedback filter means (60) to produce a predecision signal; decision means (15) for producing a decision output from the predecision symbol from the output of said subtractor; and a feedback filter (60, 60') for shifting an input decision symbol along a tapped-delay line to produce successively shifted decision symbols, weighting the successively shifted decision symbols with coefficients and combining the weighted decision symbols to produce said postcursor canceling signal at the second input of said subtractor, characterized by : a plurality of delay means (21, 22, 23, 61) for providing delays to a decision symbol to produce a short-delayed decision symbol and long-delayed decision symbol from the output of said decision means (15); and a correlation detector (28, 29, 31, 32) for detecting the correlation between one of the baseband symbols and said short-delayed and long-delayed baseband symbols, in that said feedback filter (60, 60') is arranged to receive the short-delayed decision symbol, as said input decision symbol, from said delay means if said short-delayed recovered symbol is detected by said correlation detector and receive the long-delayed decision symbol, as said input decision symbol, from said delay means if said long-delayed recovered symbol is detected by the correlation detector. The decision feedback equalizer of claim 1, characterized in that the tapped delay line of said feedforward filter (60') is divided between the first and second taps so that the first tap receives the short-delayed decision symbol and the second tap receives the long-delayed decision symbol. The decision feedback equalizer of claim 1, characterized in that a selector (70, 70') is provided for selecting one of said short-delayed and long-delayed recovered symbols having highest correlation magnitude and supplying one of said short-delayed and long-delayed decision symbols to said feedback filter (60) corresponding to the selected recovered symbol. The decision feedback equalizer of claim 1, 2 or 3, characterized in that said sensor means comprises an array of spaced sensors (10 1 -10 N , 11 1 -11 N ), further comprising: decision error detector means (19, 20) connected to said decision means (15) for detecting a decision error contained in the decision symbol; a plurality of first update circuits (40 1 -40 N ) for producing a plurality of weight coefficients from the baseband symbols of said sensors (10 1 -10 N , 11 1 -11 N ) and the decision error of said decision error detector (19, 20) and supplying said weight coefficients to said forward filter means (1) as said first coefficients; and a plurality of second update circuits (50 1 -50 M ) for producing a plurality of tap-weight coefficients from said successively shifted decision symbols of said feedback filter means (60) and the decision error of said decision error detector (19, 20) and applying the tap-weight coefficients to said multipliers (17) of the feedback filter means as said second coefficients. A method of canceling multipath components of a transmitted symbol, comprising the steps of: receiving transmitted signals and recovering baseband symbols from the received signals; forward-filtering the recovered baseband symbols by weighting the symbols with coefficients so that precursor multipath components of a transmitted symbol are canceled and combining the weighted baseband symbols to produce a forward-filtered signal; subtracting a postcursor canceling signal from the forward-filtered signal; producing a decision symbol from the subtracted forward-filtered signal; and feedback-filtering an input symbol by shifting the input symbol along a tapped-delay line to produce successively shifted decision symbols, weighting the successively shifted decision symbols and combining the weighted decision symbols to produce said postcursor canceling signal, characterized by the steps of: a) producing short-delayed and long-delayed decision symbols; b) determining whether a short-delayed recovered symbol or a long-delayed recovered symbol is detected; c) feedback-filtering the short-delayed decision symbol, as said input decision symbol, if said short-delayed recovered symbol is detected; and d) feedback-filtering the long-delayed decision symbol, as said input decision symbol, if said long-delayed recovered symbol is detected.

Inventors: TSUJIMOTO ICHIRO, TSUJIMOTO, ICHIRO, Tsujimoto, Ichiro, c/o NEC Corp.
Applicants: NIPPON ELECTRIC CO, NEC CORPORATION
Application year: 1995
Main classification: H04L25
IPC codes: H03H17, H04B1, H04L25, H04B7, H04B3