Inverter control

Abstract: An inverter control apparatus for controlling a command frequency (f1) and a command voltage (V1) to be supplied to an inverter (2) which then supplies AC power of a desired frequency determined by the command frequency and the command voltage to an induction motor (1). The inverter control apparatus receives a requested rotational speed (fref) of the motor and an actual rotational speed (f') obtained by detecting a frequency of a fluctuation component of a drive current supplied from the inverter (2) to the induction motor (1). A difference between the requested rotational speed and the actual rotational speed represents a slip amount, which is added to the requested rotational speed to produce a command frequency. The command voltage is set to a voltage when the monitored drive current is at a minimum value.

Description: The present invention relates to the control of an inverter for driving an induction motor at variable speeds by varying the drive frequency of said motor, and relates particularly to an inverter control apparatus for air conditioning systems using a heat pump cycle to provide heating and cooling. In general, compressors used in air conditioning systems for the home, small office, or small retail establishment are driven by induction motors. The slip frequency, i.e., the difference between the drive frequency and rotational frequency, of the induction motor tends to fluctuate under load, and motor efficiency thus varies. To compensate for this, vector control methods whereby the load is calculated from the motor current to operate the motor at high efficiency have been proposed as a replacement for the conventional voltage-frequency (V/F) control method obtaining an operating voltage proportional to the operating command frequency. An example of high efficiency control applying vector control technologies is shown in Fig. 1 based on the "quieting technique for inverter scroll package air conditioners" described in "Reitoshi" (a Japanese journal on refrigeration), June 1992, pp. 50 - 55. The AC power supply 6 for the induction motor 1 driving the compressor is first converted to DC power by the diode bridge 5 and smoothing capacitor 4. The DC power voltage and frequency are then adjusted by the three-phase PWM inverter 2 to change and control the speed of the induction motor 1. It is often difficult to install a rotation detector to air conditioning compressors. To detect the motor speed and drive the motor with high efficiency therefore requires that at least two current levels be detected for the induction motor 1 to detect the motor speed by calculation using an induction motor constant. The excitation current and torque current calculator 101 detects motor currents Iu and Iv using current sensors 3u and 3v to obtain the excitation current I1d and torque current I1q, respectively. The detected torque current I1q is roughly proportional to the slip frequency fs, which can therefore be calculated using the torque current I1q and the motor constant. The command frequency can then be changed to compensate the slip frequency fs, and thereby maintain the motor speed at a controlled constant level. Thus, the slip frequency is calculated by the slip frequency control means 7, and the motor speed is corrected so that the induction motor 1 rotates at a level equal to the rotational command frequency fref output by the heat pump cycle control means. Referring to Fig. 1, the adder 8 adds the rotational command frequency fref from the heat pump cycle control means and the value of a correction frequency to obtain the actual command frequency f1 for the three-phase PWM inverter 2. Identifying the primary resistance that is one motor constant is accomplished by the primary resistance identifier 102 to obtain a more precise primary resistance value based on the difference between the excitation current command I1*d obtained by the current minimizing control means 103 and the detected excitation current I1d. The actual primary voltage is then determined by the primary voltage compensator 104 based on the primary resistance thus obtained, the excitation current command I1*d, the torque current I1q, and the actual command frequency, and supplied to the three-phase PWM inverter 2. The current supply is thus controlled based on the rotational command frequency fref from the heat pump cycle control means to the minimum current level required to keep the induction motor 1 operating at the command speed. By thus minimizing the current supply, efficiency improves approximately 10% at a 1.1 kgm motor torque; improvement has also been confirmed at other torque levels. The method of the above prior art controls motor efficiency by obtaining the slip frequency from the torque current, an operation which requires a motor constant. Possible motor constants include the primary resistance, secondary resistance, and various inductances. These constants are known, however, to fluctuate greatly according to the temperature and other factors, and this fluctuation degrades the control performance, such as undesirable change in the rotational speed. Furthermore, minimising the motor current does not always maximise motor efficiency. As shown in Figure 24, when the voltage V1 applied to the motor increases, power consumption drops to the lowest level V1 (Pmin) before the motor current drops to the lowest level V1(Imin). EP 605328, a further example of the prior art, discloses a method of supplying optimal power for an alternating current electric motor driven by a static frequency converter by setting parameters which are applied to the controls of the converter and by noting the direction of variation around an extreme corresponding to minimum power absorbed by the load on the basis of one or more monitoring parameters. Therefore, an object of the present invention is to provide an inverter control apparatus for maintaining power consumption constantly at the lowest possible level in response to changes in the operating environment without requiring a motor constant input, without requiring a motor speed detector, and while compensating the slip ratio of the induction motor to maintain the motor speed at a target level. Scroll compressors recently developed for air conditioning systems are characterised by extremely low fluctuation in the gas compression torque per revolution, thereby enabling smooth, quiet operation. Current fluctuation is according small in the induction motors used to drive these scroll compressors. A further object of the present invention is therefore to provide an inverter control apparatus comprising various types of rotational period detectors as a means of maintaining compatibility with induction motors used with such scroll compressors. A further object of the present invention is to provide an inverter control apparatus for protecting the semiconductor elements used thereby without reducing the required output even under relatively exceptional operating conditions. These "exceptional conditions" include, with respect to air conditioning applications, heating even when the environmental temperature is relatively high. This is because when the environmental temperature is high, the refrigerant pressure rises, and the torque load of the compressor motor rises. For reference, the gas compression torque characteristics of a scroll compressor and rolling piston compressor to which the present invention is applicable are shown in Fig. 2, and the current wave forms of an induction motor driving these compressors are shown in Fig. 3. The graph in Fig. 2 is taken from Fig. 2 of "Low pressure type scroll compressors for air conditioners," Sawai, et al., National Technical Report, Vol. 35, No. 6, Dec. 1989, pp. 80 - 81. Virtually no gas compression torque is required at a crank angle from 0° to 90° with a rolling piston compressor, but an extremely high torque is required at a crank angle from 180° to 270°. When an induction motor is driven under such load conditions, the current consumed at crank angles requiring virtually no torque is low, but increases as the crank angle requires increasing torque. The current flowing through the motor therefore fluctuates according to the rotational angle as shown in Fig. 3, and the period of this fluctuation matches the rotational frequency. With a scroll compressor, however, the fluctuation in the gas compression torque is slight at approximately 10% of the mean, and the current fluctuation is therefore also small. To achieve the aforementioned objects, an inverter control apparatus according to one preferred embodiment of the present invention comprises: means for receiving a requested rotational speed of said motor; current detection means for detecting a drive current supplied to said motor; discrimination means for discriminating a frequency of a fluctuation component of the drive current detected by said current detection means, said fluctuation component frequency representing an actual rotational speed of said motor; slip detection means for detecting a slip amount of said induction motor by taking a difference between said actual rotational speed and said request rotational speed; command frequency producing means for producing a data commensurate with said command frequency by adding said slip amount to said request rotational speed; and command voltage generating means for generating said command voltage by monitoring said drive current and for setting said command voltage to a voltage acquired when said monitored drive current is at a minimum value. An inverter control apparatus according to another preferred embodiment of the present invention comprises: current detection means for detecting a drive current supplied to said motor; means for calculating a power consumption of the motor based on the sum of products of an integer multiple of the cycle period of the inverter output voltage and the detected drive current; and command voltage generating means for generating said command voltage by monitoring said power and for setting said command voltage to a voltage acquired when said monitored power is at a minimum value. The apparatus of this embodiment further comprises: means for receiving a requested rotational speed of said motor; current conversion means for calculating an excitation current and a torque current of the induction motor based on a phase difference between the inverter output voltage and the detected drive current; slip amount calculation means for calculating a slip amount based on said excitation current, said torque current, and a time constant of the motor; and command frequency producing means for producing a data commensurate with said command frequency by adding said slip amount to said request rotational speed. An inverter control apparatus according to yet another preferred embodiment of the present invention comprises: current detection means for detecting a drive current supplied to said motor; means for calculating a power consumption of the motor based on the sum of products of the inverter output voltage and the detected drive current; selecting means for selecting either one of said detected drive current and said power, said detected drive current being selected when said detected drive current is greater than a predefined current level, and said power being selected when said detected drive current is less than said predefined current level; and command voltage generating means for generating said command voltage by monitoring a selected data by said selecting means and for setting said command voltage to a voltage acquired when said monitored data is at a minimum value. With an inverter control apparatus according to the present invention, a current fluctuation corresponding to the actual motor speed occurs as a result of the pulsating load variation occurring when driving the compressor. The slip frequency can be measured by discriminating the frequency of this current fluctuation, and the current can therefore be minimized while controlling the slip frequency by adjusting the output voltage of the inverter. With an inverter control apparatus according to the present invention, the frequency component corresponding to the actual motor speed is lower than the drive frequency. Therefore, by attenuating the high frequency information, the drive frequency and actual motor speed component can be distinguished even though the pulsating load variation is small, and it is thus possible to minimize the current while controlling the slip frequency as in the first embodiment above. By digitizing the detected current according to the present invention, the information concerning the actual motor speed contained in the detected current data becomes the variation in the period of the digitization information. By frequency discriminating the fluctuation of this cycle, the drive frequency information and the actual motor speed component can be discriminated even when the pulsating load fluctuation is small. It is therefore possible, as with the first embodiment above, to control the slip frequency, and to minimize the current. The effects of a pulsating load change on the motor appear in the same frequency component even in the DC component of the inverter. Thus, by detecting the current of the DC component of the inverter to accomplish current minimisation control by the present invention, minimising the current during load conditions also effectively minimises power consumption. It is also possible to interpolate and correct the overall characteristics from the minimisation control result based on the partial actual command frequency by pre-storing the frequency-voltage characteristics under known standard conditions even when the operation command from the heat pump cycle control means is only a partial frequency. It is therefore possible by the present invention described above to roughly predict the conditions for achieving high efficiency across the entire operating range based on partial operating conditions. Power consumption by the motor can also be controlled to the lowest level required by the present invention described above by calculating the actual motor power consumption and adjusting the applied voltage by obtaining the sum of products of the motor current and applied voltage over an integer multiple of the cycle period of the inverter output voltage. Unless the motor operation command change suddenly, the current of each wire to a three-phase motor is a sine wave with the phase difference of each wire current offset from the others by 2/3π, and the applied voltage is also essentially a sine wave. It is thereby possible to separate the excitation current component and the torque current component by detecting the phase difference of the voltage and the current. Because the slip frequency is proportional to the torque current component and the excitation current component, compensation control of the slip frequency can be achieved by calculating the slip frequency, and motor power consumption can be minimized while maintaining the same motor speed by the present invention described above. By the present invention, high efficiency operation control can be achieved by means of power consumption minimization control when the motor current is below a predetermined level, and when the motor current rises above a particular level, current minimization control operates again to achieve priority current reduction control, thus avoiding overloading the power consumption control element. By the present invention, the voltage drop in the power consumption control element can be calculated even when the motor current increases and the voltage drop component of the power consumption control element increases, thereby using a more accurate voltage wave and achieving more accurate power consumption minimization control. The present invention will become more fully understood from the detailed description given below and the accompanying diagrams wherein: Fig. 1 is a block diagram of an inverter control apparatus according to the prior art; Fig. 2 is a gas compression torque characteristics diagram of a scroll compressor and rolling piston compressor to which the present invention is applicable; Figs. 3A and 3B are current wave diagrams of a scroll compressor and rolling piston compressor, respectively, to which the present invention is applicable; Fig. 4 is a block diagram showing an inverter control apparatus according to a first embodiment of the present invention; Fig. 5 is a block diagram showing a detail of a frequency discriminator shown in Fig. 4; Fig. 6 is a graph showing an operation characteristics of the apparatus shown in Fig. 4; Fig. 7 is a block diagram showing an inverter control apparatus according to a second embodiment of the present invention; Fig. 8 is a graph showing an operation characteristics of the apparatus shown in Fig. 7; Fig. 9 is a block diagram showing an inverter control apparatus according to a third embodiment of the present invention; Fig. 10 is a block diagram showing an inverter control apparatus according to a fourth embodiment of the present invention; Figs. 11, 12, 13 and 14 are graphs showing operations of the apparatus shown in Fig. 10; Fig. 15 is a block diagram of a control system of the invention achieved by a computer; Fig. 16 is a flow chart used to describe the operation of the computer shown in Fig. 15, particularly of the current minimization controller; Fig. 17 is a graph showing data V0 carried in a standard frequency-voltage conversion look-up table, and a modified frequency-voltage conversion look-up table; Fig. 18 is a flow chart used to describe the operation of a computer according to the fifth embodiment of the present invention; Figs. 19A and 19B are diagrams used to describe the operation of the fifth embodiment of the present invention, particularly showing the modified frequency-voltage conversion look-up table; Figs. 20A and 20B are wave forms used to describe the operation of the first or second embodiment of the present invention; Figs. 21A and 21B are wave forms used to describe the operation of the third embodiment of the present invention; Fig. 22 is a wave form used to describe the operation of the fourth embodiment of the present invention; Fig. 23 is a block diagram of a sixth embodiment of the present invention; Fig. 24 is a graph showing the relationship between the voltage applied to the motor, power consumption, and motor current; Fig. 25 is a graph showing a current-voltage characteristics when the power control device is ON in the sixth embodiment; and Fig. 26 is a flow chart is a flow chart used to describe the operation of the sixth embodiment. Embodiment 1 The first embodiment of an inverter control apparatus according to the present invention is described below with reference to Fig. 4. The inverter control apparatus is used, for example, in an air conditioning system for heating and cooling. The inverter control apparatus comprises an induction motor 1 driving the compressor used to compress and liquefy the coolant, an AC power supply 6, and a rectification circuit comprising a diode bridge 5 and smoothing capacitor 4 for converting the AC power to a DC supply. The DC supply is then converted to a three-phase AC supply by a three-phase PWM inverter 2 to drive the induction motor 1 at the desired speed. The three-phase PWM inverter 2 is a transistor-type inverter comprising an insulated gate bipolar transistor (IGBT) for controlling the drive voltage and drive frequency supplied to the induction motor 1 by varying the command voltage V1 and command frequency f1. A detail of the three-phase PWM inverter 2 is disclosed, for example, in a Japanese book "POWER ELECTRONICS NYUMON (basics)" of Sakae MURAYAMA et al. published 1994, March 25 by OHM Sha Ltd. The inverter control apparatus further comprises a current sensor 3 for producing a voltage signal Vi indicative of the current fed through one of the drive wires 1 w , an analog-to-digital converter (A/D) 14, a frequency discriminator 15, a slip frequency controller 7, an amplitude detector 18, a current minimization controller 11, an adder 8, a standard frequency-voltage conversion look-up table 9, and a multiplier 10. The rotational command frequency fref, which is a target rotational speed (rps), and the output Vi of the current sensor 3, are input to the inverter control system, which outputs the command voltage V1 and command frequency f1. The current sensor 3 produces the voltage signal Vi representing the current flowing though one of the drive wires 1 u , 1 v and 1 w , such as 1 w , of the induction motor 1. The signal Vi is applied through the amplitude detctor 18 to the current minimization controller 11, and also through the A/D converter 14 to the frequency discriminator 15. The frequency discriminator 15 extracts the frequency component corresponding to the motor speed (rps), and produces a signal f' (Fig. 20A) representing the actual rotational speed (rps) of the induction motor 1. The signal f' is applied to the slip frequency controller 7 and the current minimization controller 11. The slip frequency controller 7 is formed by a subtractor for subtracting the actual motor speed f' from the target rotational speed fref to produce a slippage fs. The adder 8 adds the target rotational speed fref and the slippage fs to produce a modified target rotational speed f1/P which can be given by the following equation (1): f1/P = fref + fs in which P represents the number of poles in the induction motor, and f1 is the command frequency of the three-phase AC voltage to be supplied to the induction motor 1. The multiplier 10 multiplies the modified target rotational speed f1/P by the number P of poles to produce the command frequency f1. The standard frequency-voltage conversion look-up table 9 is a memory for storing a standard data such as shown by a dotted line in Fig. 17, so that when the modified target rotational speed f1/P (such as f30 in Fig. 17) is input a corresponding voltage V0 (such as V30) is produced. Instead of receiving the modified target rotational speed f1/P, standard frequency-voltage conversion look-up table 9 can receive the command frequency f1. In this case the values in the table will be changed. The current minimization controller 11 receives an absolute value of the amplitude of the signal Vi from the amplitude detector 18, and monitors the signal Vi and adjusts the voltage V0 to a command voltage V1 so as to maintain a constant actual frequency. This is done by maintaining the signal Vi to the minimum value so as to maintain the current flowing through the wires 1 u , 1 v and 1 w to the minimum value. By so doing, it is possible to control the induction motor current to the smallest required level while holding the induction motor speed to the target frequency fref. A detail of the current minimization controller 11 will be described below in connection with Figs. 16 and 17. Fig. 16 is a flow chart showing the operation of the current minimization controller 11. The flow chart of Fig. 16 is carried out after selecting the command frequency (f1) or the rotational frequency (f1/P). It is assumed that the rotational frequency (f1/P) is now selected as f30 so that the standard frequency-voltage conversion look-up table 9 produces voltage V30, as understood from the dotted line in Fig. 17. At step 200, current minimization controller 11 receives the voltage V0 (in this case V30) from the standard frequency-voltage conversion look-up table 9 and also the absolute value of current signal Vi from the amplitude detector 18, and the voltage V30 is stored as V OLD and the voltage Vi is stored as Vi OLD . At step 201, the voltage V30 as stored in V OLD is subtracted by a predetermined small amount ΔV, and the difference V OLD -ΔV is stored as V NEW . At step 202, it is detected whether or not a difference between the target frequency fref and the rotational frequency f' is less than a predetermined small amount Δf, i.e., fref-f'< Δf. If not, meaning that induction motor 1 is still accelerating and not stabilized, step 202 is repeated. If yes, meaning that the induction motor 1 is stabilized, the operation goes to step 203. It is also possible to experimentally determine the time required for slip frequency compensation control to stabilize, and use a constant wait time slightly longer than this experimentally determined time. At step 203, signal Vi representing the current flowing though wire 1 w as obtained after stabilizing the induction motor 1 is input and stored as Vi NEW . At step 204, it is detected whether the absolute value of the Vi NEW is smaller than that of Vi OLD , or not. When |Vi NEW |<|Vi OLD |, it is understood that the Vi representing the current flowing through wire I w is going down. In this case, the operation goes to step 206. On the other hand, when |Vi NEW |>|Vi OLD |, it is understood that the Vi is going up. In this case the operation goes to step 205. At step 205, the sign of ΔV is changed from (+) to (-) or vice versa. At step 206, the V NEW is stored as V OLD , and Vi NEW is stored as Vi OLD . Thereafter, the operation returns to step 201. Therefore, when the operation cycle of steps 201-206 repeats a number of times with the result of step 204 being always "YES", it is understood that Vi is gradually going down. Thereafter, if the next operation cycle results in producing "NO" at step 204, it is understood that Vi has hit the minimum value and start to go up. After obtaining the minimum value of Vi in the above described manner, the operation cycles will result in producing alternately between "YES" and "NO" at step 204 to keep the signal Vi at the minimum value. The data V OLD obtained at step 206 is taken out and used as the command voltage V1 which is applied to the 3-phase PWM inverter 2. It is thereby possible to drive the induction motor 1 at the lowest possible current leaving current fluctuation equivalent to the small voltage ΔV. In the above example, since the rotational frequency (f1/P) is selected as f30, the voltage V30 (generally voltage V0) as obtained from standard frequency-voltage conversion look-up table 9 is changed to V31 (command voltage V1). In this manner, command voltages for various sampling frequencies can be obtained by storing the voltage corresponding to the minimum current obtained at the selected sampling frequencies. Such results can be stored in a memory as a modified frequency-voltage conversion look-up table. With the use of the modified frequency-voltage conversion look-up table, it is possible to quickly restore the optimum voltage at that frequency even after the command frequency fref varies temporarily to a different frequency. Referring next to Fig. 5, a block diagram of one example of the frequency discriminator 15 is shown. The input information to the frequency discriminator 15 is input to the plural bandpass filters 61, 62, ... 63. The center frequency of each of these bandpass filters 61, 62, ... 63 is different; the output from each bandpass filter is input to the corresponding amplitude detector 71, 72, ... 73 for amplitude detection. Each of the amplitude detectors 71, 72, ... 73 outputs to the maximum value detector 80, which determines which of the bandpass filter outputs is greatest. The result is then output from the maximum value detector 80 as the actual motor speed f'. It is to be noted that while a configuration comprising plural bandpass filters is shown in Fig. 5, these bandpass filters can also be achieved with digital signal processing technologies by means of time sharing, and the maximum value detector can be easily achieved in software. Using the apparatus thus described, the current flowing through the induction motor 1 when driving a rolling piston type compressor varies greatly according to the rotational angle as shown in Fig. 3, and the cycle of the fundamental wave matches the rotational cycle. Fig. 6 is a graph showing the relationship between the command frequency f1 and the actual motor speed f'. As stated above, P is the number of poles the motor has, and f1/P is the rotational speed or frequency (rps) when the motor is under a perfect no-load state. An actual motor, however, has a certain inherent load, and drives at a slightly low actual rotational frequency f'. The current flowing through the motor is a signal comprising the frequency component of the command frequency f1, and the frequency component of the actual rotational frequency f'. These two frequencies will always be different frequencies insofar as P > 1, and can therefore be easily separated. Note that with most motors, P = 2 or P = 4. An example of the motor current wave is shown in Fig. 20A assuming there are two poles (P = 2) and the motor is driven by drive frequency f1. As will be known from this wave form, the actual rotational frequency f' component is added to the fundamental period of the control frequency f1. The slip frequency controller 7 and adder 8 described above constitute the command frequency control means. The standard frequency-voltage conversion look-up table 9 and current minimization controller 11 constitute the command voltage control means. The command frequency control means, the command voltage control means, the frequency discriminator 15, and the transistor control circuit components of the three-phase PWM inverter 2 can be achieved by means of a single microprocessor. As shown in Fig. 15, such a microprocessor comprises a central processing unit (CPU) 301 for executing logical and mathematical operations; a read-only memory (ROM) 302 for storing data and commands for controlling the CPU 301; random access memory (RAM) 303 for storing the results of operations; a timer 304 for requesting external pulse frequency measurement, or CPU interrupt processing at a constant period; an A/D converter 305 for inputting external analog signals; a three-phase PWM inverter control unit 306; and a data bus 307 for connecting these components on a time-share basis. The inverter control unit 306 is a power transistor functioning as the power switching control element of the inverter, or generating a timing pulse (U, V, W and the inverted outputs U , V , W in Fig. 15) to control switching of the IGBT or other device Through normal operation of the CPU 301, ROM 302, and RAM 303 of the above microprocessor, it is possible to achieve the A/D converter 14 (Fig. 4) corresponding to the A/D converter 305 in Fig. 15, and with the timer 304 enabling interrupt processing at a regular period, the bandpass filters of the frequency discriminator 15 (Fig. 4) can be achieved by means of digital signal processing technologies. Embodiment 2 Fig. 7 is a block diagram showing an inverter control apparatus according to the second embodiment of the present invention. This embodiment differs from the first embodiment of Fig. 4 in that a low pass filter 12 and amplitude adjuster 13 are disposed between the current sensor 3 and frequency discriminator 15. The inverter control apparatus according to this embodiment can be applied to air conditioning systems using scroll compressors subject to little current fluctuation. As shown in Fig. 20A, the wave detected by the current sensor 3 comprises both the drive command frequency component f1 and the actual rotational frequency component f'. Compared with that of a rolling piston type compressor, the scroll type compressor has relatively weak rotational frequency component. The detected current is input to the low pass filter 12. The low pass filter 12 attenuates both the carrier signal component from the three-phase PWM inverter 2, and the motor drive frequency component f1. Fig. 20B is a graph of the signal wave after passing through the low pass filter 12, which results in a reduction in the relatively high drive frequency f1 component. The output from the low pass filter 12 is input to the amplitude adjuster 13 for amplitude adjustment based on the output V0 from standard frequency-voltage conversion look-up table 9. The output of the amplitude adjuster 13 assures a certain signal amplitude, and is output through the A/D converter 14 to the frequency discriminator 15 for detection of the actual rotational frequency. Note that the operating principle of the embodiment shown in Fig. 7 is identical to that shown in Fig. 4, and further description thereof is thus omitted below. Fig. 8 is a graph showing the relationship between the frequency of the detected current and the low pass filter in Fig. 7. The drive frequency component f1 and the frequency component f' obtained from actual motor operation are obtained as frequency components of size a and b, respectively, in the output from the current sensor 3; note that the size of component b is smaller than a. By passing the signal through the low pass filter 12, however, the size of the drive frequency component f1 is reduced from a to a', and the attenuation of f' is less than that of f1. The size difference between these two signal components thus disappears, and by adjusting the size of the signal containing both signal components, detection by means of a simple frequency discriminator is enabled. Embodiment 3 Fig. 9 is a block diagram showing an inverter control apparatus according to the third embodiment of the present invention. This differs from that shown in Fig. 4 in that a low pass filter 12, digitization means 21, and inverse cycle counter 22 are disposed between the current sensor 3 and frequency discriminator 23. The current wave of the induction motor 1 detected by the current sensor 3 is input to the low pass filter 12 whereby the motor drive frequency component is attenuated, and a signal as shown in Fig. 21A is produced. The current information from the low pass filter 12 is then input to the digitization means 21, which is achieved, for example, by means of an amplifier with an extremely high amplification rate with thresholds Th1 and Th2. The digitization means 21 thus produces a digitized signal as shown in Fig. 21B, and the digitized signal is then input to the inverse cycle counter 22. The inverse cycle counter 22 measures the inversion cycle periods of the digitized data. If the inversion cycle is constant, the result is also constant, and if the inversion cycle is variable, the variance also appears in the measured result. Fig. 21A is a graph used to illustrate this principle. If the digitization threshold value supplies a constant hysteresis width at the rise and fall, there is little delay until a change occurs when the amplitude is high, but when the amplitude is low, there is a great delay before a change occurs. This is shown in Fig. 21B. If inversion cycles T1, T2, T3,... in Fig. 21B are counted, the change in the inversion cycle is the actual rotational frequency f'. In other words, at the end of period T1, a signal proportional to period T1 is produced, at the end of period T2, a signal proportional to period T2 is produced, and so on. The output of the inversion cycle counter 22 is input to the frequency discriminator 23 for discrimination of the fluctuation frequency. The discriminated frequency information reflects the actual motor speed. Embodiment 4 Fig. 10 is a block diagram showing an inverter control apparatus according to the fourth embodiment of the present invention. This embodiment differs from that shown in Fig. 7 in that the current detector is disposed at the input side of the inverter circuit rather than at the output side. More specifically, a low current detection resistor 16 is inserted to the input circuit of the three-phase PWM inverter 2, and a voltage across the resistor 16 is input to an insulated amplifier 17. The output of the amplifier 17 is applied to the low pass filter 12 and also to current minimization controller 11. The characteristic operation of this embodiment is described below. Fig. 11 is a wave form diagram of the current flowing through the induction motor 1. Each phase of this three-phase AC current U, V, W is offset 1/3π from the other phases. The current flowing into the current detection resistor 16, however, is the absolute value of the peak of the three-phase current flowing through the induction motor 1 after conversion to AC current by the three-phase PWM inverter 2. This is shown in Fig. 12. In Fig. 12, the absolute value of the phase V current is greatest in the 1/3π period from 0 where time is the phase angle, and phases U and W are opposite signs, thus consuming more current than the V phase. Because the V phase current is flowing from the DC component, the current of the DC component in Fig. 12 is -V during the period from 0 to 1/3π. Similarly, the current is phase U from 1/3π to 2/3π, - W from 2/3π to π, V from π to 3/4π, -U from 4/3π to 5/4π, and W from 5/3π to 2π, thus yielding a wave as shown in Fig. 13. The current flowing through the induction motor 1 is thus a pulse wave with a cycle six times that of the drive frequency. The amplitude is thus attenuated to 0.066987 (=0.5 - sqrt(3)/4) times the amplitude of the drive frequency. The amplitude of the fundamental frequency of this pulse wave is even lower, and is 0.005456 (= 6/35π) times the amplitude of the drive frequency. Current movement due to compressor torque variation give influence to DC component, and thus the frequency and magnitude of the current movement still exist in the DC component. Fig. 22 is the current wave of the DC component. A small pulsating signal with a cycle 1/6 that of the drive frequency f1 fluctuates at the period of the rotational frequency 1/f'. If the compressor is a rolling piston type, these pulsations are large, but the pulsations are relatively small with scroll type compressors. This relationship is shown on the frequency axis of the power spectrum shown in Fig. 14. In Fig. 14, the frequency of drive frequency f1 detected by the DC component is 6 * f1, and the amplitude c is attenuated to approximately 1/18 the amplitude a of the AC component. The component f' of current fluctuation due to a pulsating torque has the same frequency and amplitude b , the two signal frequencies are separated, and the current fluctuation component due to a pulsating torque is relatively large. The attenuation of the drive frequency component obtained by the low pass filter 12 is thus 36 times using a secondary filter, and six times using the simplest primary filter, and a relatively low precision A/D converter 14 can be used downstream. In addition, protective resistors are often used with common inverter circuits to protect the transistors and other elements used to directly control the electronic circuitry. The current detection resistor 16 of this embodiment can also be used for such protection purposes, and the present embodiment of the invention can therefore be achieved without further increasing the number of parts. Note that the embodiment shown in Fig. 10 has been described as measuring the current by inserting a resistor to the DC component. The invention shall not be limited to using a resistor as the measuring means, however, and a current sensor 3 as shown in Fig. 7 can, of course, be alternatively used. Embodiment 5 The fifth embodiment of the present invention is described below. This invention relates to the storage means 9 (RAM) for storing the standard and modified frequency-voltage conversion look-up tables for producing the command voltage V1, and can be used in any of the first through fourth embodiments of the present invention. Fig. 17 is a graph in which the plotted points along the dotted line show data stored in the standard frequency-voltage conversion look-up table, and the plotted points along the solid line show data stored in the modified frequency-voltage conversion look-up table. It is noted that both tables can be stored in block (RAM) 9 in any of the embodiments disclosed herein, and in this case the command voltage V1 can be directly supplied from block 9 to the three-phase PWM inverter 2. The standard frequency-voltage conversion look-up table is previously stored in RAM 9 before the air conditioning system is supplied to the user, but the modified frequency-voltage conversion look-up table is generated and stored after setting the air conditioning system by the operation of Fig. 16 carried out for different frequencies, because the data differs by the setting condition. An example of the data stored in the standard frequency-voltage conversion look-up table is shown by data points (f10, V10) , (f20, V20), (f30, V30), and (f40, V40) along dotted line V0 in Fig. 17. Then, for obtaining data for the modified frequency-voltage conversion look-up table, first the operation frequency fref is selected. Then, the operation, for example the flow chart shown in Fig. 16, starts under particular environmental conditions as set to determine the command voltage corresponding to the selected operation frequency. For example, in Fig. 17, if operating frequency f10 is selected, the command voltage whereby the current or voltage is minimized becomes V11, thus obtaining the first data (f10, V11) for the modified frequency-voltage conversion look-up table. As the number of operating frequencies increases, the closer the optimum frequency-voltage conversion data are approached. Note that because the frequency and voltage are generally simply related, obtaining several points yielding the ideal frequency-voltage characteristic will also mean that all points between those ideal points will also yield a nearly ideal characteristic. In addition, there will be only a slight drop in efficiency even at the start of operation if the initial operating values are set to obtain the frequency-voltage characteristic under standard load conditions; if operating time is also considered, no inconvenience will occur. Fig. 18 is a flow chart of the control program carried out in the current minimization controller 11 to obtain the modified frequency-voltage conversion data which is not in the look-up table, but between the sampled points. This program is achieved by the microprocessor shown in Fig 15. When the rotational command frequency fref changes, the sampled value for the voltage V1 at that time is obtained. This process starts by inputting the rotational command frequency fref at step 210. At step 211 it is determined whether the sampled value of voltage V1 for this rotational command frequency fref is in the modified frequency-voltage conversion look-up table 9, or not. If it is in the modified frequency-voltage conversion look-up table, control passes to step 216; if not, to step 212. At step 212, a sampled frequency fp which is greater than and closest to the input frequency fref is searched for, and the voltage Vp corresponding to that sampled frequency fp is looked-up in the modified frequency-voltage conversion look-up table. If there is no such a voltage Vp corresponding to the sampled frequency fp in the modified frequency-voltage conversion look-up table, control passes to step 214 where the voltage Vp corresponding to the sampled frequency fp is looked-up in the standard frequency-voltage conversion look-up table, or a simple calculation is carried out to find a corresponding voltage Vp along the dotted line V0 in Fig. 17. In step 212, if there is a corresponding voltage Vp in the modified frequency-voltage conversion look-up table, the voltage Vp is read and the control passes to step 213. At step 213, a sampled frequency fq which is smaller than and closest to the input frequency fref is searched for, and the voltage Vq corresponding to that sampled frequency fq is looked-up in the modified frequency-voltage conversion look-up table. If there is no such a voltage Vq corresponding to the sampled frequency fq in the modified frequency-voltage conversion look-up table, control passes to step 214 where the voltage Vq corresponding to the sampled frequency fq is obtained in the same manner as described above. In step 213, if there is a corresponding voltage Vq in the modified frequency-voltage conversion look-up table, the voltage Vq is read and the control passes to step 215. At step 215, the voltage V corresponding to the command frequency fref is calculated from the current command frequency fref and the two data (fp, Vp) and (fq, Vq) according to equation (2) below based on an internal relation-ship. V = V p -V q f p -f q · f ref + f p V q -f q V p f p -f q Fig. 19A shows an example of the modified frequency-voltage conversion look-up table in which a voltage corresponding to the frequency f40 is not given. When the input frequency fref is f40, the operation in Fig. 18 is carried out such that Vp is V51 and Vq is V31. Since f40 is at the center between f30 and f40, V=0.5(V31+V51) is obtained and stored in the table as shown in Fig. 19B. Embodiment 6 Fig. 23 is a block diagram showing an inverter control apparatus according to the sixth embodiment of the present invention. According to this embodiment, the signal Vi obtained from the current sensor 3 is input to a product/sum operator 116, a zero cross timing detector 113, and the amplitude detector 112. The amplitude detector 112 detects the amplitude of the sine wave signal Vi input thereto. The zero cross timing detector 113 detects the timing at which the sine wave signal Vi crosses the zero level. Zero cross timing detection can be achieved in a hardware system inputting the output of a comparator or other detection circuit to the clock circuit, and can also be achieved in software by determining whether the read input signal is the same or opposite sign as the previous value. The output of the zero cross timing detector 113 is input to the phase comparator 122. The phase information of the primary frequency information f1 is also input from a zero cross timing detector 114 to the phase comparator 122 as the zero cross timing for phase comparison, and the phase difference is detected based on the time difference between the two timing parameters. Phase comparison result  is input to function generators 118 and 119. Function generators 118 and 119 execute sin() and cos () operations, and output to multipliers 120 and 121, respectively. Multipliers 120 and 121 multiply the supplied sine and cosine values by the output from the amplitude detector 112, and output the resulting products It and Im to the slip frequency controller 7. The slip frequency controller 7 calculates the frequency fs whereby the slip frequency will become equal to the target command frequency fref, and outputs the calculated frequency f1/P to the multiplier 10 from which the frequency command f1 is produced and applied to the three-phase PWM inverter 2. The output from the amplitude detector 112 is also input to the voltage drop compensator 115, which calculates the voltage drop component of the IGBT from the applied voltage V1, and calculates the estimated actual applied voltage. In other words, based on the voltage-current characteristics of the power control device, the voltage drop compensator 115 compensates the voltage drop component of the power control device obtained from the motor current to determine the amplitude of the voltage wave. The compensated applied voltage is input to the product/sum operator 116, which calculates the sum of products with the output from the current sensor 3, i.e., calculates the power. The power calculated by the product/sum operator 116 and the current calculated by the amplitude detector 112 are input to the selector 117, which selects one of the inputs' for output to the minimization controller 11. Switching control of the selector 117 uses the amplitude of the motor current signal Vi. Thus, when the amplitude of signal Vi is lower than a predetermined level, the power information is selected from product/sum operator 116; when the amplitude of signal Vi is higher than the predetermined level, the current amplitude is selected from sensor 3. The minimization controller 11 adjusts the applied voltage V1 to the induction motor 1 so that the power obtained as the sum of products of the voltage and the amplitude or current of the current quantity obtained by the current sensor 3 is lowest. The operation of the minimization controller 11 is described below. The actual speed of the induction motor 1 is thus maintained at the target command frequency fref, and the motor current or power can be controlled to the lowest possible levels while retaining the same output. The operation of the inverter control apparatus thus described is described in detail below. The method of detecting the slip frequency based on the phase difference between the motor current and the applied voltage is described below. According to Japanese Patent Publication S63-34718 (1988-34718), the torque current component It and excitation current component Im are calculated from the currents Iu, Iv, Iw of the three wires driving a three-phase motor as shown in equation (3) below. Im = -b·Ia + a·Ib It = a·Ia+b·Ib wherein Ia = k(Iu - 1 2 Iv - 1 2 Iw) Ib = k(- 3 2 Iv + √3 2 Iw) in which a and b are the voltages of the two phase voltage converted from the three-phase voltage, and k is a constant. Therefore, if the frequency is constant, Iu, Iv, and Iw are the same signal with the phase thereof shifted 2/3π. It is therefore possible to calculate the other current values using the information for any one current. The torque current component It and excitation current component Im are therefore related to the primary current amplitude I as shown in equation (4). Im = I * sin() It = I * cos() where  is the phase difference of the two-phase converted voltage and current. This phase difference is the same as the phase difference of the three-phase voltage and current. Therefore, the torque current component It and excitation current component Im can be obtained by obtaining the current-voltage phase difference. The slip frequency fs can thus also be obtained using the following equation (5) according to Japanese Patent Publication S63-34718 (1988-34718). fs = It 2π·T2 {Im/ (1+PT2)} where P is a differential operator, and T2 is a secondary time constant of the motor. However, because the effect of the differential operator can be ignored when there is no frequency change, the slip frequency fs can be calculated by dividing the torque current component It by the product of the excitation current component Im, secondary time constant T2, and 2π. The operation of the minimization controller 11 is described next. Fig. 26 is a control flow chart showing the operating principle of the minimization controller 11. Referring to Fig. 26, minimization control starts after selecting the command frequency fref. This minimization control starts by outputting the difference of a small voltage ΔV subtracted from the previous output voltage V OLD as the inverter output voltage (step 201) . The procedure then pauses until slip frequency compensation control stabilizes at this output voltage (step 202). Whether slip frequency compensation control has stabilized, or not, is determined by comparing the command frequency fref with the actual rotational frequency f' or slip frequency calculated from the torque current It, or other parameters. It is understood that control has stabilized when these two frequencies match. It is also possible to experimentally determine the time required for slip frequency compensation control to stabilize, and use a constant wait time slightly longer than this experimentally determined time. Once slip frequency compensation control has stabilized, a signal A representing the power being supplied to the induction motor 1 is determined and stored as variable A NEW (step 203). Instead of the signal A, the signal Vi representing the current i can be used. The values of the present variable A NEW and the variable A OLD supplied in the previous control cycle are then compared (step 204). If the present value is smaller than the previous value, control proceeds to step 206; if greater, the polarity of the low voltage ΔV is reversed (step 205), and control then proceeds to step 206. The current voltage V1 NEW and variable A NEW are then stored as the old voltage V OLD and old variable A OLD (step 206) in preparation for the next cycle, and control loops back to step 201. It is thereby possible to drive the induction motor 1 at the lowest possible current or power level leaving current or power fluctuation equivalent to the small voltage ΔV value. Furthermore, by storing the voltage corresponding to the minimum current or minimum power level obtained by this method as the optimum voltage at that frequency, it is possible to quickly restore the optimum voltage at that frequency even after the command frequency fref varies temporarily to a different frequency. It is to be noted that the current information used by the minimization controller 11 is for alternating current in this embodiment, and the effective value or maximum value, for example, is therefore used. The amplitude detector 112 shown in Fig. 23 corresponds to the means for using these values. The operation of the voltage drop compensator 115 is described next. Fig. 25 is a graph showing the current-voltage characteristics when the power control device IGBT is ON (this device is described by Takanashi, et al., in a Japanese language document "Third generation intelligent power modules," Mitsubishi Electric Technical Review, Vol. 67, No. 9, 1993, pp. 71, Fig. 11). Specifically, when the collector current Ic reaches 10 A or greater, it is known that the ON voltage Vce(sat) is increasing in a substantially linear manner. Therefore, Vce(sat) is determined by determining the collector current Ic. Because the collector current Ic corresponds to the motor current I1, if the motor current I1 is known, then Vce(sat) is known. This relationship is not environmentally dependent, but is fixed once the IGBT and PWM modulation frequency is known. These values can therefore be easily obtained by predetermining and storing the values in a microprocessor or other means for quick retrieval. It will be obvious that the present invention shall not be limited to applications with air conditioning systems, and can be applied to a wide range of apparatuses, including other air compressors producing compressed air, as well as power machinery and tools, comprising an inverter to which the command frequency f1 and command voltage V1 are input. By means of the invention thus described, it is possible to achieve high precision, high efficiency inverter control whereby power consumption can be constantly maintained at the lowest required level relative to the operating environment even with a three-phase motor by providing only one current detector and without requiring input of a motor constant. The second through fourth embodiments of the present invention described above are particularly effective with low pulsating torque loads, as are encountered with scroll compressors. It is therefore possible to achieve a high efficiency air conditioning system characterized by smooth, quiet, low vibration, low noise operation. By accumulating the most efficient frequency-voltage characteristics obtained by searching in the modified frequency-voltage conversion look-up table, it is possible by means of the fifth embodiment of the present invention described above to achieve a simple, high precision inverter comprising a learning function whereby the optimum frequency-voltage characteristics can be gradually accumulated across the entire operating range. By means of the sixth embodiment of the present invention as described above, only a single current detector is required even with a three-phase motor, rather than the two current detection means required by the prior art. It is therefore possible to achieve high precision, high efficiency inverter control whereby power consumption can be constantly maintained at the lowest required level relative to the operating environment. Furthermore, by combining current minimization control, the maximum capacitance of the power control device can be reduced, and the apparatus itself can therefore be reduced in size. The invention being thus described, it will be obvious that the same may be varied in many ways. Such variations are not to be regarded as a departure from the spirit and scope of the invention, and all such modifications as would be obvious to one skilled in the art are intended to be included within the scope of the following claims.

Claims: An inverter control apparatus for controlling an inverter (2) which, upon receipt of a command frequency (f1) and a command voltage (V1), supplies AC power of a desired frequency to an induction motor (1), said inverter control apparatus comprising: means (8) for receiving a requested rotational speed (fref) of said motor; current detection means (3: 16, 17) for detecting a drive current supplied to said motor; discrimination means (15) for discriminating a frequency of a fluctuation component of the drive current detected by said current detection means, said fluctuation component frequency representing an actual rotational speed (f') of said motor (1); slip detection means (7) for detecting a slip amount (fs) of said induction motor by taking a difference between said actual rotational speed (f') and said request rotational speed (fref); command frequency producing means (8, 10) for producing a data commensurate with said command frequency (fl) by adding said slip amount (fs) to said request rotational speed (fref); and command voltage generating means (11) for generating said command voltage (V1) by monitoring said drive current and for setting said command voltage (V1) to a voltage acquired when said monitored drive current is at a minimum value. An inverter control apparatus according to Claim 1, wherein said discriminating means (15) comprises a filter (12, 61, 62, 63) for filtering said fluctuation component. An inverter control apparatus according to Claim 2, wherein said discrimination means (15) comprises a plurality of bandpass filters (61, 62, 63) and an amplitude detection means (71, 72, 73). An inverter control apparatus according to any one of the preceding claims, wherein said discriminating means (15) comprises an amplifier (13) of which the degree of amplification is inversely proportional to the applied voltage. An inverter control apparatus according to any one of the preceding claims, wherein said discrimination means (15) comprises an analog-to-digital converter (14) for converting said detected drive current. An inverter control apparatus according to any one of the preceding claims, wherein said discrimination means (15) comprises: a filter (12) for filtering said fluctuation component; a digitizing means (21) for digitizing the filtered signal; and a detecting means (15') for detecting a frequency of the output from the digitizing means. An inverter control apparatus according to any one of the preceding claims, wherein said current detection means comprises: a DC current detection means for detecting a DC current applied to said inverter; and said discrimination means (15) for discriminating a frequency of a fluctuation component of the DC current detected by said DC current detection means. An inverter control apparatus according to any one of the preceding claims, wherein said discrimination means (15), slip detection means (7), command frequency producing means (8, 10), and command voltage generating means (11) are formed by a digital computer. An inverter control apparatus according to any one of the preceding claims, wherein said command voltage generating means (11) comprises: memory means (9) for storing a standard frequency-voltage conversion data for converting said data commensurate with said command frequency to a voltage signal, said voltage signal being adjusted to produce said command voltage. An inverter control apparatus according to Claim 9, wherein said command voltage generating means (11) is a computer based processor for adjusting said voltage signal at a real time operation. An inverter control apparatus for controlling an inverter (2) which, upon receipt of a command frequency (fl) and a command voltage (V1), supplies AC power of a desired frequency to an induction motor (1), said inverter control apparatus comprising: current detection means (3) for detecting a drive current supplied to said motor; means (116) for calculating, a power consumption of the motor based on the sum of products of the inverter output voltage and the detected drive current over an integer multiple of the cycle period of the inverter output voltage; and command voltage generating means (11) for generating said command voltage (V1) by monitoring said power and for setting said command voltage (V1) to a voltage acquired when said monitored power is at a minimum value. An inverter control apparatus for controlling an inverter (2) which, upon receipt of a command frequency (fl) and a command voltage (V1), supplies AC power of a desired frequency to an induction motor (1), said inverter control apparatus comprising: current detection means (3) for detecting a drive current supplied to said motor; means (116) for calculating a power consumption of the motor based on the sum of products of the inverter output voltage and the detected drive current; selecting means (117) for selecting either one of said detected drive current and said power, said detected drive current being selected when said detected drive current is greater than a predefined current level, and said power being selected when said detected drive current is less than said predefined current level; and command voltage generating means (11) for generating said command voltage (V1) by monitoring a selected data by said selecting means and for setting said command voltage (V1) to a voltage acquired when said monitored data is at a minimum value. An inverter control apparatus according to claim 11 or 12, further comprising: means (8) for receiving a requested rotational speed (fref) of said motor; current conversion means (118-122) for calculating an excitation current and a torque current of the induction motor based on a phase difference between the inverter output voltage and the detected drive current; slip amount calculation means (7) for calculating a slip amount (fs) based on said excitation current, said torque current, and a time constant of the motor; and command frequency producing means (8, 10) for producing a data commensurate with said command frequency (fl) by adding said slip amount (fs) to said request rotational speed (fref). An inverter control apparatus according to Claim 13, wherein said current conversion means (118-122) comprises first zero-cross detection means (113) for detecting a zero-cross point of said drive current, and a second zero-cross detection means (114) for detecting a zero-cross point of said inverter output voltage. An inverter control apparatus according to Claim 13, wherein said, power calculation means (116), command voltage generating means (11), current conversion means (118-122), slip amount calculation means (7), and command frequency producing means (8, 10) are formed by a digital computer. An inverter control apparatus according to any one of claims 11 to 15, wherein a difference of a no-load inverter output voltage subtracted by a voltage proportional to said detected drive current is substantially equal to said inverter output voltage used in said power calculation means.

Inventors: DOYAMA YOSHIAKI, ITO YOSHITERU, MATSUI KEIZO, OGATA HIDEO, WATAKABE SHUSAKU, YOSHIDA IZUMI, YOSHIOKA KANEHARU, DOYAMA, YOSHIAKI, ITO, YOSHITERU, MATSUI, KEIZO, OGATA, HIDEO, WATAKABE, SHUSAKU, YOSHIDA, IZUMI, YOSHIOKA, KANEHARU
Applicants: MATSUSHITA ELECTRIC IND CO LTD, MATSUSHITA REFRIGERATION, MATSUSHITA ELECTRIC INDUSTRIAL CO., LTD., MATSUSHITA REFRIGERATION COMPANY
Application year: 1995
Main classification: H02P5
IPC codes: H02P23, G01P3