OBJECT POSITION DETECTOR WITH EDGE MOTION FEATURE

Abstract: A proximity sensor system includes a sensor matrix array having a characteristic capacitance on horizontal and vertical conductors connected to sensor pads. The capacitance changes as a function of the proximity of an object or objects to the sensor matrix. The change in capacitance of each node in both the X and Y directions of the matrix due to the approach of an object is converted to a set of voltages in the X and Y directions. These voltages are processed by circuitry to develop electrial signals representative of the centroid of the profile of the object, i.e., its position in the X and Y dimensions. Noise reduction and background level setting techniques inherently available in the architecture are employed. The speed of the cursor movement depends on the one of the display it resides.

Description: SE££IEI£ΔIIΩE Title of The Invention Object Position Detector With Edge Motion Feature Background Of The Invention 1. Field Of The Invention The present invention relates to object position sensing transducers and systems. More particularly, the present invention relates to object position recognition useful in applications such as cursor movement for computing devices and other applications, and especially to cursor movement with enhanced edge-motion features. 2. The Prior Art Numerous devices are available or have been proposed for use as object position detectors for use in computer systems and other applications. The most familiar of such devices is the computer "mouse". While extremely popular as a position indicating device, a mouse has mechanical parts and requires a surface upon which to roll its position ball. Furthermore, a mouse usually needs to be moved over long distances for reasonable resolution. Finally, a mouse requires d e user to lift a hand from the keyboard to make the cursor movement, thereby upsetting the prime purpose, which is usually typing on the computer. Trackball devices are similar to mouse devices. A major difference, however is that, unlike a mouse device, a trackball device does not require a surface across which it must be rolled. Trackball devices are still expensive, have moving parts, and require a relatively heavy touch as do die mouse devices. They are also large in size and doe not fit well in a volume- sensitive application like a laptop computer. There are several available touch-sense technologies which may be employed fo use as a position indicator. Resistive-membrane position sensors are known and used in severa applications. However, they generally suffer from poor resolution, the sensor surface is expose to die user and is thus subject to wear. In addition, resistive-membrane touch sensors ar relatively expensive. A one-surface approach requires a user to be grounded to die sensor fo reliable operation. This cannot be guaranteed in portable computers. An example of a one-surfac approach is the UnMouse product by MicroTouch, of Wilmington, MA A two-surface approac has poorer resolution and potentially will wear out very quickly in time. Resistive tablets are taught by United States Patent No. 4,680,430 to Yoshikawa United States Patent No. 3,497,617 to Ellis and many others. The drawback of all suc approaches is the high power consumption and the high cost of the resistive membrane employed. Surface Acoustic Wave (SAW) devices have potential use as position indicators However, this sensor technology is expensive and is not sensitive to light touch. In addition SAW devices are sensitive to residue buildup on the touch surfaces and generally have poo resolution. Strain gauge or pressure plate approaches are an interesting position sensin technology, but suffer from several drawbacks. This approach may employ piezo-electri transducers. One drawback is that the piezo phenomena is an AC phenomena and may b sensitive to the user's rate of movement. In addition, strain gauge or pressure plate approaches ar somewhat expensive because special sensors are required. Optical approaches are also possible but are somewhat limited for several reasons All would require light generation which will require external components and increase cost an power drain. For example, a "finger-breaking" infra-red matrix position detector consumes hig power and suffers from relatively poor resolution. There have been numerous attempts to provide a device for sensing die position o a thumb or other finger for use as a pointing device to replace a mouse or trackball. Desirabl attributes of such a device are low power, low profile, high resolution, low cost, fast response and ability to operate reliably when die finger carries electrical noise, or when die touch surface i contaminated with dirt or moisture. Because of the drawbacks of resistive devices, many attempts have been made t provide pointing capability based on capacitively sensing me position of die finger. United States Patent No. 3,921,166 to Volpe teaches a capacitive matrix in which the finger changes die transcapacitance between row and column electrodes. United States Patent No. 4,103,252 to Bobick employs four oscillating signals to interpolate x and y positions between four capacitive electrodes. United States Patent No.4,455,452 to Schuyler teaches a capacitive tablet wherein die finger attenuates die capacitive coupling between electrodes. United States Patent No. 4,550,221 to Mabusth teaches a capacitive tablet wherein the effective capacitance to "virtual ground" is measured by an oscillating signal. Each row or column is polled sequentially, and a rudimentary form of interpolation is applied to resolve the position between two rows or columns. An attempt is made to address the problem of electrical interference by averaging over many cycles of the oscillating waveform. The problem of contamination is addressed by sensing when no finger was present, and applying a periodic calibration during such no-finger-present periods. United States Patent No. 4,639,720 to Rympalski teaches a tablet for sensing the position of a stylus. The stylus alters the transcapacitance coupling between row and column electrodes, which are scanned sequentially. United States Patent No. 4,736,191 to Matzke teaches a radial electrode arrangement under the space bar of a keyboard, to be activated by touching with a thumb. This patent teaches the use of total touch capacitance, as an indication of the touch pressure, to control die velocity of cursor motion. Pulsed sequential polling is employed to address die effects of electrical interference. United States Patent Nos. 4,686,332 and 5,149,919, to Greanias, teaches a stylus and finger detection system meant to be mounted on a CRT. As a finger detection system, it's X/Y sensor matrix is used to locate die two matrix wires carrying d e maximum signal. Witii a coding scheme these two wires uniquely determine the location of d e finger position to die resolution of die wire stepping. For stylus detection, Greanias first coarsely locates it, tiien develops a virtual dipole by driving all lines on one side of die object in one direction and all lines on die opposite side in the opposite direction. This is done d ree times with different dipole phases and signal polarities. Assuming a predetermined matrix response to d e object, the three measurements present a set of simultaneous equations tiiat can be solved for position. United States Patent No. 4,733,222 to Evans is d e first to teach a capacitance touch measurement system tiiat interpolates to a high degree. Evans teaches a three terminal measurement system that uses a drive, sense and electrode signal set (3 signals) in its matrix, and bases the measurement on d e attenuation effect of a finger on the electrode node signal (uses a capacitive divider phenomena). Evans sequentially scans dirough each drive set to measure the capacitance. From the three largest responses an interpolation routine is applied to determine finger position. Evans also teaches a zeroing technique that allows "no-finger" levels to be cancelled out as part of d e measurement. United States Patent No. 5,016,008 to Gruaz describes a touch sensitive pad tiiat also uses interpolation. Gruaz uses a drive and sense signal set (2 signals) in d e touch matrix and like Evans relies on the attenuation effect of a finger to modulate the drive signal. The touch matrix is sequentially scanned to read die response of each matrix line. An interpolation program tiien selects d e two largest adjacent signals in both dimensions to determine die finger location, and ratiometrically determines me effective position from those 4 numbers. Gerpheide, PCT application US90/04584, publication No. W091/03039, United States Patent No. 5,305,017 applies to a touch pad system a variation of the virtual dipole approach of Greanias. Gerpheide teaches the application of an oscillating potential of a given frequency and phase to all electrodes on one side of the virtual dipole.and an oscillating potential of die same frequency and opposite phase to those on the other side. Electronic circuits develop a "balance signal" which is zero when no finger is present, and which has one polarity if a finger is on one side of die center of me virtual dipole, and die opposite polarity if die finger is on die opposite side. To acquire the position of the finger initially, the virtual dipole is scanned sequentially across the tablet. Once die finger is located, it is "tracked" by moving the virtual dipole toward me finger once me finger has moved more than one row or column. Because die virtual dipole method operates by generating a balance signal that is zero when the capacitance does not vary widi distance, it only senses the perimeter of die finger contact area, rather tiian me entire contact area. Because die method relies on synchronous detection of d e exciting signal, it must average for long periods to reject electrical interference, and hence it is slow. The averaging time required by this me od, togetiier wid die necessity to search sequentially for a new finger contact once a previous contact is lost, makes this method, like tiiose before it, fall short of die requirements for a fast pointing device mat is not affected by electrical interference. It should also be noted that all previous touch pad inventions mat used interpolation placed rigorous design requirements on dieir sensing pad. Greanias and Evans use a complicated and expensive drive, sense and electrode line scheme to develop their signal. Gruaz and Gerpheide use a two signal drive and sense set. In die present invention die driving and sensing is done on d e same line. This allows die row and column sections to be symmetric and equivalent This in turn allows independent calibration of all signal paths, which makes board layout simpler and less constraining, and allows for more unique sensor topologies. The shortcomings of the inventions and techniques described in d e prior art can also be traced to d e use of only one set of driving and sensing electronics, which was multiplexed sequentially over d e electrodes in d e tablet. This arrangement was cost effective in die days of discrete components, and avoided offset and scale differences among circuits. The sequential scanning approach of previous systems also made them more susceptible to noise. Noise levels could change between successive measurements, thus changing the measured signal and the assumptions used in interpolation routines. Finally, all previous approaches assumed a particular signal response for finger position versus matrix position. Because the transfer curve is very sensitive to many parameters and is not a smooth linear curve as Greanias and Gerpheide assume, such approaches are limited in me amount of interpolation tiiey can perform. In prior co-pending application serial No. 08/115,743, filed August 31, 1993, now United States Patent No. , a two-dimensional capacitive sensing system equipped widi a separate set of drive/sense electronics for each row and for each column of a capacitive tablet is disclosed. All row electrodes are sensed simultaneously, and all column electrodes are sensed simultaneously. The sensed signals are processed by analog circuitry. It is thus an object of the present invention to provide a two-dimensional capacitive sensing system equipped widi a separate set of drive/sense electronics for each row and for each column of a capacitive tablet, wherein all row electrodes are sensed simultaneously, and all column electrodes are sensed simultaneously. It is a further object of die present invention to provide an electronic system tiiat is sensitive to the entire area of contact of a finger or otiier conductive object widi a capacitive tablet, and to provide as output the coordinates of some measure of die center of this contact area while remaining insensitive to me characteristic profile of die object being detected. It is a further object of the present invention to provide an electronic system mat provides as output some measure of area of contact of a finger or other conductive object widi a capacitive tablet Yet another object of die present invention is to provide a two-dimensional capacitive sensing system equipped widi a separate set of drive/sense electronics for each row and for each column of a capacitive tablet, wherein all row electrodes are sensed simultaneously, and all column electrodes are sensed simultaneously and wherein die information defining die location of a finger or other conductive object is processed in digital form. It is a further object of die present invention to provide a two-dimensional capacitive sensing system wherein all row electrodes are sensed simultaneously, and all column electrodes are sensed simultaneously and wherein die location of a finger or otiier conductive object within a peripheral region of a sensing plane can optionally cause cursor "edge motion" on display screen allowing control of large cursor excursions from a small sensing plane with a single gesture. Brief Description Of The Invention With die advent of very high levels of integration, it has become possible to integrate many channels of driving/sensing electronics into one integrated circuit, along widi di control logic for operating diem, and the interface electronics to allow die pointing device t communicate directly widi a host microprocessor. The present invention uses adaptive analog techniques to overcome offset and scale differences between channels, and can thus sense either transcapacitance or self-capacitance of all tablet rows or columns in parallel. This parallel-sensing capability, made possible by providing one set of electronics per row or column, allows di sensing cycle to be extremely short, tiius allowing fast response while still maintaining immunity to very high levels of electrical interference. The present invention comprises a position-sensing technology particularly useful for applications where finger position information is needed, such as in computer "mouse" o trackball environments. However the position-sensing technology of the present invention has much more general application man a computer mouse, because its sensor can detect and report if one or more points are being touched. In addition, die detector can sense the pressure of the touch. According to a preferred embodiment of the present invention, referred to herein as a "finger pointer" embodiment, a position sensing system includes a position sensing transducer comprising a touch-sensitive surface disposed on a substrate, such as a printed circuit board, including a matrix of conductive lines. A first set of conductive lines runs in a first direction and is insulated from a second set of conductive lines running in a second direction generally perpendicular to d e first direction. An insulating layer is disposed over die first and second sets of conductive lines. The insulating layer is thin enough to promote significant capacitive coupling between a finger placed on its surface and the first and second sets of conductive lines. Sensing electronics respond to die proximity of a finger, conductive object, or an object of high dielectric constant (i.e., greater than about 5) to translate die capacitance changes of die conductors caused by object proximity into digital information which is processed to derive position and touch pressure information. Its output is a simple X, Y and pressure value of die one object on its surface. In all descriptions herein, fingers are to be considered interchangeable widi conductive objects and objects of high dielectric constant Different prior art pad scan techniques have different advantages in different environments. Parallel drive/sense techniques according to the present invention allow input samples to be taken simultaneously, thus all channels are affected by the same phase of an interfering electrical signal, greatiy simplifying die signal processing and noise filtering. There are two drive/sense methods employed in the touch sensing technology of die present invention. According to a first and presendy preferred embodiment of the invention, die voltages on all of the X lines of the sensor matrix are simultaneously moved, while die voltages of die Y lines are held at a constant voltage, widi die complete set of sampled points simultaneously giving a profile of the finger in die X dimension. Next, die voltages on all of the Y lines of the sensor matrix are simultaneously moved, while the voltages of the X lines are held at a constant voltage to obtain a complete set of sampled points simultaneously giving a profile of the finger in the ot er dimension. According to a second drive sense metiiod, die voltages on all of die X lines of die sensor matrix are simultaneously moved in a positive direction, while die voltages of the Y lines are moved in a negative direction. Next, d e voltages on all of the X lines of me sensor matrix are simultaneously moved in a negative direction, while die voltages of die Y lines are moved in a positive direction. This technique doubles die effect of any transcapacitance between die two dimensions, or conversely, halves the effect of any parasitic capacitance to ground. In both methods, the capacitive information from die sensing process provides a profile of d e proximity of e finger to d e sensor in each dimension. As presendy preferred, both embodiments men take tiiese profiles and derive a digital value representing the centroid for X and Y position and derive a second digital value for d e Z pressure information. The digital information may be direcdy used by a host computer. Analog processing of the capacitive information may also be used according to the present invention. The position sensor of these embodiments can only report d e position of one object on its sensor surface. If more than one object is present, die position sensor of this embodiment computes die centroid position of die combined set of objects. However, unlike prior art, because die entire pad is being profiled, enough information is available to discern simple multi-finger gestures to allow for a more powerful user interface. According to another aspect of me present invention, several power reduction techniques which can shut down die circuit between measurements have been integrated into d e system. This is possible because the parallel measurement technique according to die presen invention is so much faster than prior art techniques. According to a furtiier aspect of the invention, a variety of noise reductio techniques are integrated into d e system. According to yet another aspect of the present invention, a capacitance measurement technique which is easier to calibrate and implement is employed. According to another aspect of die present invention, when me presence of a finger or other conductive object is sensed within a defined peripheral region of the sensing plane, d e control of cursor motion may be changed to provide "edge motion" to allow control of large curso excursions on a display screen from a single gesture executed on a small sensing plane. Brief Description Of The Drawings Figure 1 is an overall block diagram of the capacitive position sensing system of the present invention. Figure 2a is a top view of an object position sensor transducer according to a presendy preferred embodiment of the invention showing the object position sensor surface layer including a top conductive trace layer and conductive pads connected to a bottom trace layer. Figure 2b is a bottom view of die object position sensor transducer of FIG. 2a showing the bottom conductive trace layer. Figure 2c is a composite view of the object position sensor transducer of FIGS. 2a and 2b showing both die top and bottom conductive trace layers. Figure 2d is a cross-sectional view of die object position sensor transducer of FIGS. 2a-2c. Figure 3 is a block diagram of sensor decoding electronics which may be used widi e sensor transducer in accordance with a preferred embodiment of the present invention. Figure 4a is a simplified schematic diagram of a charge integrator circuit which may be used in die present invention. Figure 4b is an illustrative schematic diagram of the charge integrator circuit of FIG. 4a. Figure 5 is a timing diagram of the operation of charge integrator circuit of FIGS. 4a and 4b. Figure 6 is a schematic diagram of an illustrative filter and sample hold circuit for use in die present invention. Figure 7 is a more detailed block diagram of a presendy preferred arrangement of A/D converters for use in the present invention. Figure 8 is a block diagram of an illustrative arithmetic unit which may be used in die present invention. Figure 9 is a block diagram of a calibration unit which may be used widi the arithmetic unit of FIG. 8. Figure 10 is a schematic diagram of a bias voltage generating circuit useful in die present invention. Figure 11 is a diagram of die sensing plane illustrating die edge motion feature of the object position sensor of d e present invention. Figure 12 is a schematic diagram illustrating hardware implementation of d e determination of whether a finger or otiier object is present in die peripheral regions of die sensing plane. Figure 13 is a schematic diagram illustrating hardware implementation of die edge motion feature of me present invention. Figures 14a- 14c are flow diagrams illustrating a process according to d e present invention for recognizing tap and drag gestures. Detailed Description Of A Preferred Embodiment This application is a continuation-in-part of co-pending application serial No 08/ , filed September 2, 1994, which is a continuation-in-part of co-pendin application serial No. 08/115,743, filed August 31, 1993, now United States Patent No , which is a continuation-in-part of co-pending application serial No. 07/895,934, file June 8, 1992. The present invention continues die approach disclosed in d e parent application and provides more unique features not previously available. These improvements provide a mo easily integrated solution, increased sensitivity, and greater noise rejection, increased dat acquisition rate and decreased power consumption. The present invention allows for continuou self calibration to subtract out the effects of environmental changes and allows for enhanced curso control from edge motion on a sensing plane. Those of ordinary skill in die art will realize that the following description of di present invention is illustrative only and not in any way limiting. Otiier embodiments of th invention will readily suggest themselves to such skilled persons. The present invention brings together in combination a number of unique feature which allow for new applications not before possible. Because the object position sensor of di present invention has very low power requirements, it is beneficial for use in battery operated o low power applications such as lap top or portable computers . It is also a very low cost solution has no moving parts (and is therefore virtually maintenance free), and uses the existing printe circuit board traces for sensors. The sensing technology of the present invention can be integrate into a computer motherboard to even further lower its cost in computer applications. Similarly, i other applications e sensor can be part of an already existent circuit board. Because of its small size and low profile, the sensor technology of the presen invention is useful in lap top or portable applications where volume is an important consideration The sensor technology of the present invention requires circuit board space for only a singl sensor interface chip that can interface directly to a microprocessor, plus the area needed on di printed circuit board for sensing. Referring first to FIG. 1, a simplified block diagram of die capacitive positio sensing system 6 of the present invention is presented. Capacitive position sensing system 6 ca accurately determine die position of a finger 8 or other conductive object proximate to or touchin a sensing plane 10. The capacitance of a plurality of conductive lines running in a first directio (e.g., "X") is sensed by X input processing circuitry 12 and die capacitance of a plurality o conductive lines running in a second direction (e.g., "Y") is sensed by Y input processin circuitry 14. The sensed capacitance values are digitized in botii X input processing circuitry 12 and Y input processing circuitry 14. The outputs of X input processing circuitry 12 and Y input processing circuitry 14 are presented to aridimetic unit 16, which uses the digital information to derive digital information representing d e position and pressure of d e finger 8 or otiier conductive object relative to die sensing plane 10. The X, Y, and Z outputs of aridimetic unit 16 are directed to motion unit 18 which provides die cursor motion direction signals to die host computer. The X, Y, and Z outputs of aridimetic unit 16 are also directed to gesture unit 20, which is used to recognize certain finger gestures performed by a user on sensing plane 10. The virtual button output from gesture unit 20 may be used by motion unit 18 to implement some of the functionality of motion unit 18. The sensor material can be anything that allows creation of a conductive X/Y matrix of pads. This includes not only standard PC boards, but also includes but is not limited to flexible PC boards, conductive elastomer materials, silk-screened conductive lines, and piezo- electric Kynar plastic materials. This renders it useful as well in any portable equipment application or in human interface where the sensor needs to be molded to fit witiiin die hand. The sensor can be conformed to any tiiree dimensional surface. Copper can be plated in two layers on most any surface contour producing the sensor. This will allow the sensor to be adapted to die best ergonomic form needed for any particular application. This coupled widi d e "light-touch" feature will make it effortless to use in many applications. The sensor can also be used in an indirect manner, i.e it can have an insulating foam material covered by a conductive layer over die touch sensing surface and be used to detect any object (not just conductive) that presses against it's surface. Small sensor areas are practical, i.e., a presendy conceived embodiment takes about 1.5"x 1.5" of area, however those of ordinary skill in die art will recognize that die area is scaleable for different applications. The matrix area is scaleable by eitiier varying the matrix trace spacing or by varying die number of traces. Large sensor areas are practical where more information is needed. Besides simple X and Y position information, the sensor technology of die present invention also provides finger pressure information. This additional dimension of information may be used by programs to control special features such as "brush-width" modes in Paint programs, special menu accesses, etc., allowing provision of a more natural sensory input to computers. It has also been found useful for implementing "mouse click and drag" modes and for simple input gestures. The user will not even have to touch the surface to generate d e minimum reaction. This feature can gready minimize user strain and allow for more flexible use. The sense system of the present invention depends on a transducer device capable of providing position and pressure information regarding the object contacting die transducer. Referring now to FIGS. 2a-2d, top, bottom, composite, and cross-sectional views, respectively, are shown of a presently-preferred sensing plane 10 comprising a touch sensor array 22 for use in the present invention. Since capacitance is exploited by this embodiment of die present invention, die surface of touch sensor array 22 is designed to maximize the capacitive coupling to a finger or other conductive object A presently preferred touch sensor array 22 according to the present invention comprises a substrate 24 including a set of first conductive traces 26 disposed on a top surface 28 diereof and run in a first direction to comprise row positions of the array. A second set of conductive traces 30 are disposed on a bottom surface 32 thereof and run in a second direction preferably orthogonal to die first direction to form the column positions of the array. The top and bottom conductive traces 26 and 30 are alternately in contact with periodic sense pads 34 comprising enlarged areas, shown as diamonds in FIGS. 2a- 2c. While sense pads 34 are shown as diamonds in FIGS. 2a-2c, any shape, such as circles, which allows diem to be closely packed is equivalent for purposes of diis invention. As an arbitrary convention herein, the first conductive traces 26 will be referred to as being oriented in die "X" or "row" direction and may be referred to herein sometimes as "X lines" and d e second conductive traces 30 will be referred to as being oriented in die "Y" or "column" direction and may be referred to herein sometimes as 'Υ lines". The number and spacing of these sense pads 34 depends upon die resolution desired. For example, in an actual embodiment constructed according to die principles of the present invention, a 0.10 inch center- to-center diamond-shaped pattern of conductive pads disposed along a matrix of 15 rows and 15 columns of conductors is employed. Every otiier sense pad 34 in each direction in die pad pattern is connected to conductive traces on the top and bottom surfaces 28 and 32, respectively of substrate 24. Substrate 24 may be a printed circuit board, a flexible circuit board or any of a number of available circuit interconnect technology structures. Its thickness is unimportant as long as contact may be made theretiirough from the bottom conductive traces 30 to tiieir sense pads 34 on die top surface 28. The printed circuit board comprising substrate 24 can be constructed using standard industry techniques. Board ti ickness is not important. Connections from die conductive pads 34 to d e bottom traces 30 may be made employing standard plated-ti rough hole techniques well known in die printed circuit board art In an alternate embodiment of he present invention, die substrate material 24 may have a thickness on die order of 0.005 to 0.010 inches. Then d e diamonds on the upper surface 28 and d e plated through holes that connect to the lower surface traces 30, can be omitted, further reducing die cost of die system. An insulating layer 36 is disposed over die sense pads 34 on top surface 28 to insulate a human finger or other object therefrom. Insulating layer 36 is preferably a thin layer (i.e., approximately 5 roils) to keep capacitive coupling large and may comprise a material, such as mylar, chosen for its protective and ergonomic characteristics. The term "significant capacitive coupling" as used herein shall mean capacitive coupling having a magnitude greater than about 0.5 pF. There are two different capacitive effects taking place when a finger approaches the touch sensor array 22. The first capacitive effect is trans-capacitance, or coupling between sense pads 34, and die second capacitive effect is self-capacitance, or coupling to virtual ground. Sensing circuitry is coupled to die sensor array 22 of the present invention and responds to changes in eitiier or both of these capacitances. This is important because die relative sizes of the two capacitances change greatly depending on die user environment The ability of die present invention to detect changes in bom self capacitance and trans-capacitance results in a very versatile system having a wide range of applications. According to die preferred embodiment of the invention, a position sensor system including touch sensor array 22 and associated position detection circuitry will detect a finger position on a matrix of printed circuit board traces via die capacitive effect of finger proximity to me sensor array 22. The position sensor system will report the X, Y position of a finger placed near die sensor array 22 to much finer resolution than die spacing between die row and column traces 26 and 30. The position sensor according to this embodiment of the invention will also report a Z value proportional to die oudine of that finger and hence indicative of die pressure widi which die finger contacts die surface of insulating layer 36 over die sensing array 22. According to die presendy preferred embodiment of die invention, a very sensitive, light-touch detector circuit may be provided using adaptive analog and digital VLSI techniques. The circuit of die present invention is very robust and calibrates out process and systematic errors. The detector circuit of die present invention will process the capacitive input information and provide digital information which may be presented directly to a microprocessor. According to tiiis embodiment of die invention, sensing circuitry is contained on a single sensor processor integrated circuit chip. The sensor processor chip can have any number of X and Y "matrix" inputs. The number of X and Y inputs does not have to be equal. The Integrated circuit has a digital bus as output. In the illustrative example disclosed in FIGS. 2a-2d herein, the sensor array has 15 traces in both the X and Y directions. The sensor processor chip tii us has 15 X inputs and 15 Y inputs. An actual embodiment constructed according to die principles of the present invention employed 18 traces in d e X direction and 24 traces in die Y direction. Those of ordinary skill in the art will recognize that die size of die sensing matrix which may be employed in me present invention is arbitrary and will be dictated largely by design choice. The X and Y matrix nodes are driven and sensed in parallel, widi die capacitive information from each line indicating how close a finger is to that node. The scanned information provides a profile of the finger proximity in each dimension. According to tins aspect of the present invention, the profile centroid is derived in botii the X and Y directions and is the position in tiiat dimension. The profile curve of proximity is also integrated to provide the Z information. There are two drive and sense metiiods employed in the touch sensing technology of die present invention. According to a first and presendy preferred embodiment of the invention, the voltages on all of die X lines of the sensor matrix are simultaneously moved, while die voltages of die Y lines are held at a constant voltage. Next, die voltages on all of the Y lines of the sensor matrix are simultaneously moved, while the voltages of the X lines are held at a constant voltage. This scanning method accentuates die measurement of capacitance to virtual ground provided by the finger. Those of ordinary skill in the art will recognize that order of these two steps is somewhat arbitrary and may be reversed. According to a second drive sense method, die voltages on all of die X lines of die sensor matrix are simultaneously moved in a positive direction, while the voltages of die Y lines are moved in a negative direction. Next, die voltages on all of the X lines of the sensor matrix are simultaneously moved in a negative direction, while the voltages of the Y lines are moved in a positive direction. This second drive/sense method accentuates transcapacitance and de- emphasizes virtual ground capacitance. As widi the first drive/sense method, those of ordinary skill in die art will recognize that order of these two steps is somewhat arbitrary and may be reversed. Referring now to FIG. 3, a block diagram of the presently preferred sensing circuitry 40 for use according to die present invention is presented. This block diagram, and die accompanying disclosure, relates to die sensing circuitry in one dimension (X) only, and includes die X input processing circuitry 12 of FIG. 1. Those of ordinary skill in die art will appreciate mat an identical circuit would be used for sensing die opposite (Y) dimension and would include die Y input processing circuitry 14 of FIG. 1. Such skilled persons will further note tiiat die two dimensions do not need to be orthogonal to one another. For example, tiiey can be radial or of any otiier nature to match the contour of d e touch sensor array and other needs of d e system. Those of ordinary skill in die art will recognize tiiat d e technology disclosed herein could be applied as well to a one-dimensional case where only one set of conductive traces is used. The capacitance at each sensor matrix node is represented by equivalent capacitors 42-1 tiirough 42-n. The capacitance of capacitors 42-1 through 42-n comprises the capacitance of die matrix conductors and has a characteristic background value when no object (e.g., a finger) is proximate to die sensing plane of die sensor matrix. As an object approaches the sensing plane die capacitance of capacitors 42-1 tiirough 42-n increases in proportion to the size and proximity of the object. According to the present invention, the capacitance at each sensor matrix node is measured simultaneously using charge integrator circuits 44-1 through 44-n. Charge-integrator circuits 44-1 through 44-n serve to inject charge into the capacitances 42-1 through 42-n, respectively, and to develop an output voltage proportional to the capacitance sensed on die corresponding X matrix line. Thus charge-integrator circuits 44-1 through 44-n are shown as bidirectional amplifier symbols. Each charge-integrator circuit 44-1 through 44-n is supplied with an operating bias voltage by bias-voltage generating circuit 46. As used herein, the phrase "proportional to the capacitance" means that d e voltage signal generated is a monotonic function of the sensed capacitance. In d e embodiment described herein, the voltage is directly and linearly proportional to the capacitance sensed. Those of ordinary skill in d e art will recognize that other monotonic functions, including but not limited to inverse proportionality, and non-linear proportionality such as logarithmic or exponential functions, could be employed in die present invention without departing from the principles disclosed herein. In addition current-sensing as well as voltage-sensing techniques could be employed. According to a presently preferred drive/sense method used in the present invention, the capacitance measurements are performed simultaneously across all inputs in one dimension to overcome a problem which is inherent in all prior art approaches mat scan individual inputs. The problem with the prior-art approach is that it is sensitive to high frequency and large amplitude noise (large dv/dt noise) that is coupled to die circuit via the touching object Such noise may distort die finger profile because of noise appearing in a later scan cycle but not an earlier one, due to a change in die noise level. The present invention overcomes this problem by "taking a snapshot" of all inputs simultaneously in X and then Y directions (or visa versa). Because die injected noise is proportional to die finger signal strength across all inputs, it is therefore symmetric around d e finger centroid. Because it is symmetric around die finger centroid it does not affect d e finger position. Additionally, the charge amplifier performs a differential measuring function to further reject common-mode noise. Because of the nature of the charge integrator circuits 44-1 through 44-n, their outputs will be changing over time and will have die desired voltage output for only a short time. As presently preferred, filter circuits 48-1 tiirough 48-n are implemented as sample and hold switched capacitor filters. The desired voltage is captured by the filter circuits 48-1 through 48-n. As controlled by control circuitry, 56, the filter circuits 48-1 through 48-n will filter out any high frequency noise from the sensed signal. This is accomplished by choosing the capacitor for the filter to be much larger than the output capacitance of charge integrator circuits 44- 1 tiirough 44-n. In addition, those of ordinary skill in the art will recognize that die switched capacitor filter circuits 48-1 through 48-n will capture the desired voltages and store diem. According to the present invention, the capacitance information obtained in voltage form from die capacitance measurements is digitized and processed in digital format Accordingly, the voltages stored by filter circuits 48-1 through 48-n are stored in sample/hold circuits 50-1 through 50-n so that the remainder of the circuitry processes input data taken at the same time. Sample/hold circuits 50-1 through 50-n may be configured as conventional sample/hold circuits as is well known in the art. The sampled analog voltages at the outputs of sample/hold circuits 50-1 tiirough 50-n are digitized by analog-to-digital (A/D) converters 52. As presendy preferred, A D converters 52 resolve the input voltage to a 10-bit wide digital signal (a resolution of one part in 1,024), although those of ordinary skill in die art will realize that other resolutions may be employed. A/D converters 52 may be conventional successive approximation type converters as is known in the art. Given the charge integrator circuitry employed in die present invention, the background level (no object present) of die charge integrator outputs will be about 1 volt The ΔV resulting from the presence of a finger or other object will typically be about 0.4 volt The voltage range of the A/D converters 52 should merefore be in die range of between about 1-2 volts. An important consideration is the minimum and maximum voltage reference points for the A/D converters (V.^ and V max ). It has been found tiiat noise will cause position jitter if these reference voltages are fixed points. A solution to tiiis problem which is employed in die present invention is to dynamically generate die V,^ and V,,^ reference voltages from reference capacitances 42-Vmin and 42-Vmax, sensed by charge integrator circuits 44-Vmin and 44- Vmax and processed by filter circuits 48-Vmin and 48-Vmax and stored in sample hold circuits 50-Vmin and 50- Vmax . In this manner, any common mode noise present when die signals are sampled from the sensor array will also be present in the W πύn and V m8Λ reference voltage values and will tend to cancel. Those of ordinary skill in die art will realize tiiat reference capacitances 44- Vmin and 44- Vmax may eidier be discrete capacitors or extra traces in the sensor array. According to die present invention, die V,^ reference voltage is generated from a capacitor having a value equal to the lowest capacitance expected to be encountered in the sensor array with no object present (about 12pF assuming a 2 inch square sensor array). The V,,,,, reference voltage is generated from a capacitor having a value equal to the largest capacitance expected to be encountered in the sensor array with an object present (about 16pF assuming a 2 inch square sensor array). The outputs of A/D converters 52 provide inputs to arithmetic unit 16. As will be more fully disclosed widi reference to FIG. 8, die function of arithmetic unit 16 is to compute tiie weighted average of die signals on die individual sense lines in both the X and Y directions in die touch sensor array 22. Thus, arithmetic unit 16 is shared by die X input processing circuitry 12 and die Y input processing circuitry 14 as shown in FIG. 1. Control circuitry 56 of FIG. 3 orchestrates die operation of the remainder of die circuitry. Because die system is discretely sampled and pipelined in its operation, control circuitry 56 is present to manage the signal flow. The functions performed by control circuitry 56 may be conventionally developed via what is commonly known in die art as a state machine or microcontroller. The structure and operation of the individual blocks of FIG. 3 will now be disclosed. Referring now to FIGS. 4a, 4b, and 5, a typical charge integrator circuit will be described. Charge integrator circuit 44 is shown as a simplified schematic diagram in FIG. 4a and as an illustrative schematic diagram in FIG. 4b. The timing of the operation of charge integrator circuit 44 is shown in FIG. 5. These timing signals are provided by die controller block 56. Charge integrator circuit 44 is based on die fundamental physical phenomena of using a current to charge a capacitor. If die capacitor is charged for a constant time by a constant current, then a voltage will be produced on the capacitor which is inversely proportional to die capacitance. The capacitance to be charged is die sensor matrix line capacitance 42 in parallel with an internal capacitor. This internal capacitor will contain the voltage of interest Referring now to FIG. 4a, a simplified schematic diagram of an illustrative charge integrator circuit 44 is shown. A charge integrator circuit input node 60 is connected to one of die X (or Y) lines of the sensor matrix. A first shorting switch 62 is connected between die charge integrator circuit input node 60 and V D r die positive supply rail. A second shorting switch 64 is connected between die charge integrator circuit input node 60 and ground, the negative supply rail. A positive constant current source 66 is connected to V D D. the positive supply rail and to the charge integrator circuit input node 60 and through a first current source switch 68. A negative constant current source 70 is connected to ground and to die charge integrator circuit input node 60 and through a second current source switch 72. It is obvious that other high and low voltage rails could be used in place of V DD and ground. A first internal capacitor 74 is connected between V D D and output node 76 of charge integrator circuit 44. A positive voltage storage switch 78 is connected between output node 76 and input node 60. A second internal capacitor 80 has one of its plates connected to ground through a switch 82 and to output node 76 of charge integrator circuit 44 tiirough a switch 84, and die other one of its plates connected to input node 60 through a negative voltage storage switch 86 and to V DD through a switch 88. The capacitance of first and second internal capacitances 74 and 80 should be a small fraction (i.e., about 10%) of the capacitance of die individual sensor matrix lines. In a typical embodiment, the sensor matrix line capacitance will be about lOpF and die capacitance of capacitors 74 and 80 should be about lpF. According to die presendy preferred embodiment of die invention, the approach used is a differential measurement for added noise immunity, die benefit of which is that any low frequency common mode noise gets subtracted out. For the following discussion, it is to be assumed tiiat all switches are open unless they are noted as closed. First, the sensor matrix line is momentarily shorted to V DD through switch 62, switch 78 is closed connecting capacitor 74 in parallel with the capacitance of the sensor line. Then the parallel capacitor combination is discharged widi a constant current from current source 70 through switch 72 for a fixed time period. At the end of the fixed time period, switch 78 is opened, thus storing the voltage on the sensor matrix line on capacitor 74. The sensor line is then momentarily shorted to ground through switch 64, and switches 82 and 86 are closed to place capacitor 80 in parallel widi the capacitance of the sensor line. Switch 68 is closed and die parallel capacitor combination is charged widi a constant current from current source 66 for a fixed time period equal to die fixed time period of die first cycle. At die end of die fixed time period, switch 86 is opened, thus storing the voltage on die sensor matrix line on capacitor 80. The first and second measured voltages are then averaged. This is accomplished by opening switch 82 and closing switches 88 and 84, which places capacitor 80 in parallel widi capacitor 74. Because capacitors 74 and 80 have die same capacitance, die resulting voltage across diem is equal to the average of die voltages across each individually. This final result is the value mat is tiien passed on to die appropriate one of filter circuits 48-1 tiirough 48-n. The low frequency noise, notably 50/60 Hz and their harmonics, behaves as a DC current component that adds in one measurement and subtracts in the other. When die two results are added together that noise component averages to zero. The amount of noise rejection is a function of how quickly in succession the two opposing charge-up and charge-down cycles are performed as will be disclosed herein. One of die reasons for the choice of this charge integrator circuit is that it allows measurements to be taken quickly. Referring now to FIG. 4b, a more complete schematic diagram of an illustrative embodiment of charge integrator circuit 44 of the simplified diagram of FIG. 4a is shown. Input node 60 is shown connected to V DD and ground through pass gates 90 and 92, which replace switches 62 and 64 of FIG. 4a. Pass gate 90 is controlled by a signal ResetUp presented to its control input and pass gate 92 is controlled by a signal ResetDn presented to its control input Those of ordinary skill in d e art will recognize tiiat pass gates 90 and 92, as well as all of die otiier pass gates which are represented by the same symbol in FIG. 4b may be conventional CMOS pass gates as are known in the art. The convention used herein is that the pass gate will be off when its control input is held low and will be on and present a low impedance connection when its control input is held high. P-Channel MOS transistors 94 and 96 are configured as a current mirror. P- Channel MOS transistor 94 serves as die current source 66 and pass gate 98 serves as switch 68 of FIG. 4a. The control input of pass gate 98 is controlled by a signal StepUp. N-Channel MOS transistors 100 and 102 are also configured as a current mirror. N-Channel MOS transistor 100 serves as the current source 70 and pass gate 104 serves as switch 72 of FIG. 4a. The control input of pass gate 104 is controlled by a signal StepDn. P-Channel MOS transistor 106 and N-Channel MOS transistor 108 are placed in series widi P-Channel MOS current mirror transistor 96 and N-Channel MOS current mirror transistor 102. The control gate of P-Channel MOS transistor 106 is driven by an enable signal EN, which turns on P-Channel MOS transistor 106 to energize the current mirrors. This device is used as a power conservation device so tiiat die charge integrator circuit 44 may be turned off to conserve power when it is not in use. N-Channel MOS transistor 108 has its gate driven by a reference voltage Vbias, which sets the current through current mirror transistors 96 and 108. The voltage Vbias is set by a servo feedback circuit as will be disclosed in more detail widi reference to FIG. 10. Those of ordinary skill in die art will appreciate diat this embodiment allows calibration to occur in real time (via long time constant feedback) tiiereby zeroing out any long term effects due to sensor environmental changes. In a current embodiment of the invention, Vbias is common for all charge integrator circuits 44-1 through 44-n and 44- Vmax and 44-Vmin. Note that proper sizing of MOS transistors 102 and 108 may provide temperature compensation. This is accomplished by taking advantage of the fact that the threshold of N- Channel MOS transistor 108 reduces with temperature while the mobility of both N-Channel MOS transistors 102 and 108 reduce with temperature. The threshold reduction has the effect of increasing the current while the mobility reduction has the effect of decreasing the current By proper device sizing these effects can cancel each other out over a significant part of the operating range. Capacitor 74 has one plate connected to V DD and the other plate connected to the output node 76 and to the input node 60 through pass gate 110, shown as switch 78 in FIG. 4a. The control input of pass gate 110 is driven by the control signal SUp. One plate of capacitor 80 is connected to input node 60 through pass gate 112 (switch 86 in FIG. 4a) and to VDD through pass gate 114 (switch 82 in FIG.4a). The control input of pass gate 112 is driven by the control signal SDn and the control input of pass gate 114 is driven by the control signal ChUp. The otiier plate of capacitor 80 is connected to ground tiirough N-Channel MOS transistor 116 (switch 82 in FIG. 4a) and to output node 76 through pass gate 118 (switch 84 in FIG. 4a). The control input of pass gate 118 is driven by control signal Share. Referring now to FIGS.4a, 4b and die timing diagram of FIG.5, the operation of charge integrator circuit 44 during one scan cycle may be observed. First the EN (enable) control signal goes active by going to Ov. This turns on the current mirrors and energizes the charge and discharge current sources, MOS transistors 94 and 100. The ResetUp control signal is active high at this time, which shorts the input node 60 (and die sensor line to which it is connected) to V DD . The SUp control signal is also active high at this time which connects capacitor 74 and die output node 76 to input node 60. This arrangement guarantees that the following discharge portion of die operating cycle always starts from a known equilibrium state. The discharge process starts after ResetUp control signal goes inactive. The StepDn control signal goes active, connecting MOS transistor 100, the discharge current source, to d e input node 60 and its associated sensor line. StepDn is active for a set amount of time, and die negative constant current source discharges d e combined capacitance of d e sensor line and capacitor 74 tiius lowering its voltage during that time. StepDn is then turned off. A short time later the SUp control signal goes inactive, storing the measured voltage on capacitor 74. That ends die discharge cycle. Next, die ResetDn control signal becomes active and shorts the sensor line to ground. Simultaneously the SDn and ChDn control signals become active and connect capacitor 80 between ground and the sensor line. Capacitor 80 is discharged to ground, guaranteeing that e following charge up cycle always starts from a known state. The charge up cycle starts after ResetDn control signal becomes inactive and die StepUp control signal becomes active. At tiiis point the current charging source, MOS transistor 94, is connected to the sensor line and supplies a constant current to charge die sensor line by increasing die voltage tiiereon. The StepUp control signal is active for a set amount of time (preferably equal to the time for the previously mentioned cycle) allowing the capacitance to charge, and men it is turned off. The SDn control signal then goes inactive, leaving the measured voltage across capacitor 80. The averaging cycle now starts. First the voltage on capacitor 80 is level shifted. This is done by the ChDn control signal going inactive, letting one plate of d e capacitor 80 float Then die ChUp control signal goes active, connecting the second plate of die capacitor to V DD - Then die Share control signal becomes active which connects the first plate of capacitor 80 to output node 76, tiius placing capacitors 74 and 80 in parallel. This has die effect of averaging die voltages across d e two capacitors, thus subtracting out common-mode noise as previously described. This average voltage is also tiien available on output node 76. Those of ordinary skill in die art will recognize that the environmental alternating current and otiier low frequency noise-canceling feature inherent in the averaging of the voltages obtained in die discharge and charge cycles is most effective when the two cycles are performed very close togetiier in time. According to die present invention, die ChDn and ChUp signals should be asserted widi respect to each other within a time period much less than a quarter of the period of die noise to be cancelled in order to take advantage of this feature of the present invention. According to the present invention, two different drive/sense methods have been disclosed. Those of ordinary skill in the art will readily observe tiiat the charge integrator circuit 44 disclosed widi reference to FIGS. 4a, 4b, and 5 is adaptable to operate according to either scanning method disclosed herein. As is clear from an understanding of die operation of charge integrator circuit 44, its output voltage is only available for a short period of time and is subject to environmental noise. In order to minimize the effects of noise, a switched capacitor filter circuit 48 is used. Referring now to FIG. 6, a schematic diagram of an illustrative switched capacitor filter circuit 48 which may be used in the present invention is shown. Those of ordinary skill in the art will recognize this switched capacitor filter circuit, which comprises an input node 120, a pass gate 122 having a control input driven by a Sample control signal, a capacitor 124 connected between the output of die pass gate 126 and a fixed voltage such as ground, and an output node comprising the common connection between the capacitor 124 and the output of the pass gate 126. In a typical embodiment capacitor 116 will have a capacitance of about 10 pF. As will be appreciated by persons of ordinary skill in die art, d e switched capacitor filter 48 is in part a sample/hold circuit and has a filter time constant which is K times the period of sample, where K is the ratio of capacitor 124 to the sum of capacitors 74 and 80 of the charge integrator circuit 44 of FIGS. 4a and 4b to which it is connected. The switched capacitor filter circuit 48 further reduces noise injection in the system. In the preferred embodiment, K = 10/2 = 5. Those of ordinary skill in the art will recognize that otiier types of filter circuits, such as RC filters, may be employed in me present invention. Referring now to FIG. 7, a more detailed block diagram of a presendy preferred arrangement of A/D converters 52 of FIG. 3 is presented. There are fewer A/D converters than there are lines in the touch sensor array, and die inputs to die A/D converters are multiplexed to share each of the individual A/D converters among several lines in the touch sensor array. The arrangement in FIG. 7 is more efficient in the use of integrated circuit layout area tiian providing individual A/D converters for each input line. In the embodiment illustrated in FIG. 7, twenty-four conductive line traces are assumed for the sensor array 10 of FIGS. 2a-2d. As shown in FIG. 7, the outputs of sample/hold circuits 50-1 through 50-24 are fed to the analog data inputs of analog multiplexer 130. Analog multiplexer 130 has six outputs, each of which drives the input of an individual A D converter 52- 1 through 52-6. The internal arrangement of analog multiplexer 130 is such that four different ones of die inputs are multiplexed to each of die outputs. Analog multiplexer 130 has been conceptually drawn as six internal multiplexer blocks 132-1 through 132-6. In die example shown in FIG. 7, inputs taken from sample/hold circuits 50-1 tiirough 50-4 are multiplexed to die output of internal multiplexer block 132-1 which drives A/D converter 52-1. Similarly, inputs taken from sample/hold circuits 50-5 through 50-8 are multiplexed to die output of internal multiplexer block 132-2 which drives A/D converter 52-2; inputs taken from sample hold circuits 50-9 tiirough 50-12 are multiplexed to die output of internal multiplexer block 132-3 which drives A/D converter 52-3; inputs taken from sample/hold circuits 50-13 through 50-16 are multiplexed to the output of internal multiplexer block 132-4 which drives A/D converter 52-4; inputs taken from sample/hold circuits 50-17 through 50-20 are multiplexed to the output of internal multiplexer block 132-5 which drives A/D converter 52-5; and inputs taken from sample/hold circuits 50-21 through 50-24 are multiplexed to the output of internal multiplexer block 132-6 which drives A/D converter 52-6. Analog multiplexer 130 has a set of control inputs schematically represented by bus 134. In the illustrative embodiment shown in FIG. 7, each of internal multiplexers 132-1 through 132-6 are four-input multiplexers and thus control bus 134 may comprise a two-bit bus for a one- of-four selection. Those of ordinary skill in the art will recognize that the arrangement of FIG. 7 is merely one of a number of specific solutions to the task of A/D conversion from twenty-four channels, and that other satisfactory equivalent arrangements are possible. In a straightforward decoding scheme, multiplexers 132-1 through 132-6 will pass, in sequence, the analog voltages present on their first through fourth inputs on to the inputs of A D converters 52-1 through 52-6 respectively. After the analog values have settled in the inputs of A/D converters 52-1 through 52-6, a CONVERT command is asserted on common A/D control line 136 to begin the A/D conversion process. When the A/D conversion process is complete, the digital value representing the input voltage is stored in registers 138-1 tiirough 138-6. As presently preferred, registers 138-1 tiirough 138-6 may each comprise a two-word register, so that one word may be read out of the registers to arithmetic unit 54 while a second word is being written into die registers in order to maximize the speed of the system. The design of such registers is conventional in the art Referring now to FIG. 8, a more detailed block diagram of the arithmetic unit 16 is presented. Those of ordinary skill in die art will appreciate that arithmetic unit 16 processes information from both the X and Y dimensions, i.e., from X input processing circuit 12 and Y input processing circuit 14 of FIG. 1. Before disclosing the structural configuration of arithmetic unit 16, it is helpful to understand the preferred method by which the centroid position of an object proximate to the sensor array 22 is determined according to die present invention. According to a presendy preferred embodiment of d e invention, the object position in either direction may be determined by evaluating the weighted average of die capacitances measured on die individual sense line of the sensor array 10. In the following discussion, die X direction is used, but tiiose of ordinary skill in die art will recognize that die discussion applies to die determination of die weighted average in the Y direction as well. As is well known, the weighted average may be determined as follows: ∑ I X ΔCI X position = -^ [Eq. 1] ∑ ΔC i«0 where ΔC, = C, - CO,. C, is the capacitance presentiy being measured on the ith trace and CO, is the value measured on that same trace at some past time when no object was present. In terms of these past and present capacitance measurements, the position can be expressed as: ∑ i x (Cι - CO/) Xposition = -^ [Eq. 2] ∑ (Ci - CO/) . -0 Using the distributive property of multiplication over addition, this expression is seen to be equivalent to: - Σ (i x COι) + ∑ Cx C/) Xposition = —- '^-^ - Ώ [Eq. 3] - ∑ (CO + ∑ (CO i=0 i=0 where the negative terms in both the numerator and denominator are offsets and represent die background value of the capacitances with no object present. If the term 0 N is used to represent the numerator offset and die term 0 D is used to represent d e denominator offset, Eq. 3 may be re¬ written as: a -ON + ∑ d ' ) Xposition = ^ [Eq. 4] - OD + ∑ (C0 i-0 Referring now to FIG. 8, it may be seen that arithmetic unit 16 includes X numerator and denominator accumulators 150 and 152 and Y numerator and denominator accumulators 154 and 156. The source of operand data for X numerator and denominator accumulators 150 and 152 and Y numerator and denominator accumulators 154 and 156 are die registers 138-1 tiirough 138-6 in each (X and Y) direction of the sensor array 22 of FIG. 1. The X and Y denominator accumulators 152 and 156 sum up the digital results from the A D conversions. The X and Y numerator accumulators 150 and 154 compute the weighted sum of the input data ratiier tiian the straight sum. Accumulators 150, 152, 154, and 156 may be configured as hardware elements or as software running on a microprocessor as will be readily understood by tiiose of ordinary skill in die art. As may be seen from an examination of FIG. 8, numerator accumulators 150 and 154 compute the expression of Eq. 4 : ∑ x C [Eq. 5] and denominator accumulators 152 and 156 compute the expression of Eq.4: n ∑ Ci [Eq. 6] The contents of X and Y numerator and denominator offset registers 158, 160, 162, and 164 are subtracted from die results stored in die accumulators 150, 152, 154, and 156 in adders 166, 168, 170, and 172. Adder 166 subtracts the offset 0 NX stored in X numerator offset register 158. Adder 168 subtracts die offset 0 D χ stored in X denominator offset register 160. Adder 170 subtracts d e offset ONY stored in Y numerator offset register 162. Adder 172 subtracts die offset O ΓJ Y stored in Y denominator offset register 164. The numerator denominator pairs are divided by division blocks 174 and 176 to produce die X and Y position data, and die X and Y denominator pair is used by block 178 to produce Z axis (pressure) data. The function performed by block 178 will be disclosed later herein. The offsets 0 DX , 0 NX , 0 DY , and O j ^y are sampled from die accumulator contents when directed by calibration unit 180. Persons of ordinary skill in die art will readily appreciate tiiat the architecture of the system of the present invention may be distributed in a number of ways, several of which involve the availability of a microprocessor, whether it be in a host computer to which the system of the present invention is connected or somewhere between the integrated circuit described herein and a host computer. Embodiments of die present invention are contemplated wherein the accumulated numerator and denominator values representing the summation terms are delivered to such a microprocessor along with the 0 N and 0 D offset values for processing, or where all processing is accomplished by a programmed microprocessor as is known in die art Initially, the numerator and denominator accumulators 150, 152, 154, and 156 are set to zero during system startup. If the multiplexed A/D converters as shown in FIG. 7 are employed, the digitized voltage data in the first word of register 138-1 (representing the voltage at the output of sample hold circuit 50-1) is added to the sum in the accumulator and the result stored in the accumulator. In succession, the digitized voltage values stored in the first word of registers 138-2 through 138-6 (representing the voltage at the outputs of sample/hold circuits 50-5, 50-9, 50-17, and 50-21, respectively) are added to the sums in the accumulators and the results stored in the accumulators. As previously mentioned, A/D converters 52-1 through 52-6 may at this time be converting the voltages present at the outputs of sample hold circuits 50-2, 50-6, 50-10, 50-14, 50-18, and 50-22 and storing the digitized values in the second words of registers 138-1 through 138-6 respectively. Next, in succession, the digitized voltage values stored in the second words of registers 138-1 through 138-6 (representing the voltage at die outputs of-sample hold circuits 50- 2, 50-6, 50-10, 50-14, 50-18, and 50-22, respectively) are added to the sum in die accumulator and the result stored in the accumulator. Next, in succession, the digitized voltage values stored in the first words of registers 138-1 tiirough 138-6 (representing die voltage at die outputs of sample/hold circuits 50-3, 50-7, 50-11, 50-15, 50-19, and 50-23, respectively) are added to die sum in die accumulator and d e result stored in die accumulator, followed by digitized voltage values stored in the second words of registers 138-1 through 138-6 (representing the voltage at the outputs of sample/hold circuits 50-4, 50-8, 50-12, 50-16, 50-20, and 50-24, respectively). At this point in time, the accumulators hold the sums of all of the individual digitized voltage values. The digital values stored in die 0 N and 0 D offset registers 158 and 164 are now respectively subtracted from the values stored in the numerator and denominator accumulators. The division operation in dividers 174 and 176 then completes the weighted average computation. The division operation may also be performed by an external microprocessor which can fetch the values stored in die accumulators or perform the accumulations itself. As the 0 N and 0 D offset values are presently derived by an external microprocessor, the additional processing overhead presented to such external microprocessor by this division operation is minimal. Alternately, a dedicated microprocessor may be included on chip to handle these processing tasks without departing from the invention disclosed herein. The above disclosed processing takes place within about 1 millisecond and may be repeatedly performed. Current mouse standards update position information 40 times per second, and thus die apparatus of the present invention may easily be operated at this repetition rate. Because of the nature of the method employed in the present invention, an opportunity exists to provide additional noise immunity widiout requiring additional hardware in die system of die present invention. While it is apparent that after the above-disclosed sequence has been performed, the accumulators may be cleared and the process repeated, the values may also be allowed to remain in the accumulators. If this is done, an averaging function may be implemented to further filter out noise. According to tiiis aspect of the invention, a number of samples are taken and run through the accumulators without clearing them at the end of die processing sequence. As presentiy preferred, twenty-five samples are processed before a single division result is taken for use by the system, thus greatly reducing d e effects of transient system noise spikes. Those of ordinary skill in die art will recognize tiiat the number of samples taken prior to clearing the accumulators is a matter of design choice dictated by factors such as data acquisition rates, data processing rates etc. The system of the present invention is adaptable to changing conditions, such as component aging, changing capacitance due to humidity, and contamination of die touch surface, etc. In addition, the present invention effectively minimizes ambient noise. According to the present invention, these effects are taken into consideration in three ways. First, the offset values 0 N and 0 D are dynamically updated to accommodate changing conditions. Second, a servo- feedback circuit is provided to determine the bias voltage used to set d e bias of die charge- integrator circuits 44-1 through 44-n. Third, as previously disclosed herein, d e reference voltage points for V ma χ and ,^ of die A/D converters are also dynamically altered to increase die signal to noise margin. Referring now to FIG. 9, a block diagram of a calibration unit 150 which may be used with the arithmetic unit of FIG. 8 is presented. The calibration unit 150 executes an algoritiim to establish the numerator and denominator offset values by attempting to determine when no finger or other conductive object is proximate to the touch sensor array 22. As previously disclosed, the 0 N and 0 D offset values represent the baseline values of the array capacitances with no object present. These values are also updated according to the present invention since baseline levels which are too low or too high have the effect of shifting the apparent position of the object depending on the sign of the error. These values are established by selection of the values read when no object is present at the sensor array 22. Since there is no external way to "know" when no object is present at sensor array 22, an algorithm according to another aspect of the present invention is used to establish and dynamically update these offset values. When the calibration unit sees a Z value which appears typical of the Z values when no finger is present, it instructs the offset registers (158, 160, 162, and 164 of FIG. 8) to reload from the current values of the accumulators. According to a presentiy preferred embodiment of the invention, the decision to update the offset values is based on the behavior of the sensor array 22 in only one of the X or Y directions, but when the decision is made all four offsets (0 NX , OD X , ONY, and O DY ) are updated. In otiier embodiments of the invention, the decision to update may be individually made for each direction according to the criteria set forth herein. The calibration algorithm works by monitoring changes in a selected one of the denominator accumulator values. According to the present invention, it has been observed mat the sensitivity to changes in capacitance of one of the sets of conductive lines in the touch sensor array 22 is greater than the sensitivity to changes in capacitance of the other one of the sets of conductive lines in the touch sensor array 22. Experience suggests that die set of conductive lines having d e greater sensitivity to capacitance changes is die one which is physically located above d e conductive lines in the other direction and therefore closest to me touch surface of the sensor array 22. The upper set of conductive lines tends to partially shield die lower set of conductive lines from capacitive changes occurring above die surface of the sensor array 22. The finger pressure is obtained by summing the capacitances measured on die sense lines. This value is already present in the denominator accumulator after subtracting the offset 0 D . A finger is present if the pressure exceeds a suitable threshold value. This threshold may be chosen experimentally and is a function of surface material and circuit timing. The threshold may be adjusted to suit the tastes of die individual user. The pressure reported by die device is a simple function f(X D Y D ) of die denominators for die X and Y directions as implemented in block 178 of FIG. 8. Possible functions include choosing one preferred denominator value, or summing the denominators. In a presendy preferred embodiment the smaller of the two denominators is chosen. This choice has die desirable effect of causing die pressure to go below die threshold if d e finger moves slighdy off die edge of d e pad, where the X sensors are producing valid data, but die Y sensors are not, or vise versa. This acts as an electronic bezel which can take the place of a mechanical bezel at the periphery of die sensor area. In the example of FIG. 8, the Y denominator is chosen for monitoring because it is the most sensitive. The chosen denominator is referred to as Z for the purposes of the calibration algorithm. The current saved offset value for this denominator is referred to as 0 . The goal of the calibration algorithm is to track gradual variations in the resting Z level while making sure not to calibrate to the finger, nor to calibrate to instantaneous spikes arising from noise. As will be apparent to those of ordinary skill in the art from die following disclosure, die calibration algorithm could be implemented in digital or analog hardware, or in software. In a current embodiment actually tested by die inventors, it is implemented in software. As Z values arrive in the calibration unit, they are passed through filter 182. History buffer 184, which operates in conjunction with filter 182, keeps a "running average" of recent Z values. When a new Z value arrives, the current running average F z is updated according to d e formula: new F z = α(old F z ) + (1- α)Z [Eq. 7] where α is a constant factor between 0 and 1 and typically close to 1 and Z is the current Z value. In the preferred embodiment, alpha is approximately 0.95. The intention is for F z to change slowly enough to follow gradual variations, without being gready affected by short perturbations i Z. The filter 182 receives a signal ENABLE from control unit 186. The running average F z is updated based on new Z values only when ENABLE is asserted. If ENABLE is deasserted, F z remains constant and is unaffected by current Z. The history buffer 184 records the several most recent values of F z . In the present embodiment, die history buffer records die two previous F z values. The history buffer might be implemented as a shift register, circular queue, or analog delay line. When die history buffer receives a REWIND signal from control unit 186, it restores the current running average F z to the oldest saved value. It is as if die filter 182 were "retroactively" disabled for an amount of time corresponding to the depth of the history buffer. The purpose of the history buffer is to permit such retroactive disabling. The current running average F z is compared against the current Z value and the current offset C^ by absolute difference units 188 and 190, and comparator 192. Absolute difference unit 188 subtracts the values Z and F z and outputs the absolute value of their difference. Absolute difference unit 190 subtracts the values O z and F z and outputs the absolute value of their difference. Comparator 192 asserts the UPDATE signal if the output of absolute difference unit 188 is less than the output of absolute difference unit 190, i.e., if F z is closer to Z than it is to C^. The UPDATE signal will tend to be asserted when the mean value of Z shifts to a new resting level. It will tend not to be asserted when Z makes a brief excursion away from its normal resting level. The filter constant a determines the length of an excursion which will be considered "brief for this purpose. Subtractor unit 194 is a simple sub tractor that computes the signed difference between Z and O z . This subtractor is actually redundant with subtractor 172 in figure 8, and so may be merged with it in the actual implementation. The output C z of this subtractor is the calibrated Z value, an estimate of the finger pressure. This pressure value is compared against a positive and negative Z threshold by comparators 196 and 198. These thresholds are shown as ZJH and -ZTH, although they are not actually required to be equal in magnitude. If pressure signal C z is greater than Z jH , die signal FINGER is asserted indicating the possible presence of a finger. The Zrø tiireshold used by die calibration unit is similar to that used by the rest of the system to detect the presence of the finger, or it may have a different value. In the present embodiment, the calibration Z-rø is set somewhat lower than the main rø to ensure that the calibration unit makes a conservative choice about die presence of a finger. If pressure signal C z is less than - Z-ΠJ, the signal FORCE is asserted. Since O z is meant to be equal to the resting value of Z with no finger present, and a finger can only increase the sensor capacitance and tiius the value of Z, a largely negative C z implies that the device must have incorrectly calibrated itself to a finger, which has just been removed. Calibration logic 200 uses this fact to force a recalibration now that the finger is no longer present. Control logic 186 is responsible for preventing running average F z from being influenced by Z values that occur when a finger is present Output ENABLE is generally off when the FINGER signal is true, and on when die FINGER signal is false. However, when FINGER transitions from false to true, the control logic also pulses the REWIND signal. When FINGER transitions from true to false, the control logic waits a short amount of time (comparable to the depth of the history buffer) before asserting ENABLE. Thus, the running average is prevented from following Z whenever a finger is present, as well as for a short time before and after the finger is present Calibration logic 200 produces signal RECAL from the outputs of the three comparators 192, 196, and 198. When RECAL is asserted, the offset registers 0 N and Oβ will be reloaded from the current accumulator values. RECAL is produced from the following logic equation: RECAL = FORCE or (UPDATE and not FINGER). [Eq. 8] In addition, calibration logic 200 arranges to assert RECAL once when the system is first initialized, possibly after a brief period to wait for the charge integrators and other circuits to stabilize. From the descriptions of control logic 186 and calibration logic 200, it will be apparent to those of ordinary skill in die art that these blocks can be readily configured using conventional logic as a matter of simple and routine logic design. It should be obvious to any person of ordinary skill in die art that the calibration algorithm described is not specific to die particular system of charge integrators and accumulators of the current invention. Rather, it could be employed in any touch sensor which produces proximity or pressure data in which it is desired to maintain a calibration point reflecting the state of the sensor when no finger or spurious noise is present Referring now to FIG. 10, a bias voltage generating circuit 46 useful in the present invention is shown in schematic diagram form. According to a presendy preferred embodiment of die invention, all of the bias transistors 108 (FIG. 4b) of charge integrator circuits 44-1 through 44-n have their gates connected to a single source of bias voltage, although persons of ordinary skill in the art recognize that other arrangements are possible. There are a number of ways in which to generate the bias voltage required by charge integrator circuits 44-1 tiirough 4-n. As may be seen from an examination of FIG. 10, the bias voltage generating circuit 46 is an overdamped servo system. A reference source which approximates the current source function of a typical one of the charge integrator circuits 44-1 through 44-n includes a capacitor 202 having one of its plates grounded. The otiier one of its plates is connected to die V DD power supply through a first pass gate 204 and to a current source transistor 206 through a second passgate 208. A filter circuit 210, identical to the filter circuits 48-1 through 48-n and controlled by the same signal as filter circuits 48-1 through 48-n is connected to sample the voltage on capacitor 202 in the same manner that the filter-and-sample/hold circuits 48-1 through 48-n sample the voltages on the sensor conductor capacitances in the sensor array 22. The output of filter circuit 210 is fed to the non-inverting input of a weak transconductance amplifier 212, having a bias current in the range of from about 0.1-0.2μA. The inverting input of the transconductance amplifier 212 is connected to a fixed voltage of about 1 volt generated, for example, by diode 214 and resistor 216. The output of transconductance amplifier 212 is shunted by capacitor 218 and also by capacitor 220 through passgate 222. Capacitor 220 is chosen to be much larger than capacitor 218. In a typical embodiment of the present invention, capacitor 218 may be about 0.2pF and capacitor 220 may be about lOpF. Capacitor 220 is connected to the gate of N-Channel MOS transistor 224, which has its drain connected to the drain and gate of P-Channel MOS transistor 226 and its source connected to die drain and gate of N-Channel MOS transistor 228. The source of P-Channel MOS transistor 226 is connected to V DD and the source of N-Channel MOS transistor 228 is connected to ground. The common drain connection of transistors 224 and 228 is the bias voltage output node. An optional passgate 230 may be connected between a fixed voltage source (e.g., about 2 volts) and capacitor 220. Passgate 230 may be used to initialize the bias generating circuit 46 on startup by charging capacitor 220 to the fixed voltage. During each sample period, the filter circuit 210 takes a new sample. If the new sample differs from me previous sample, die output voltage of transconductance amplifier 212 will change and start to charge or discharge capacitor 218 to a new voltage. Passgate 222 is switched on for a short time (i.e., about lμsec) and the voltages on capacitors 218 and 220 try to average themselves. Due to the large size difference between capacitors 218 and 220, capacitor 218 cannot supply enough charge to equalize the voltage during die period when passgate 222 is open. This arrangement prevents large changes in bias voltage from cycle to cycle. Capacitor 202 should look as much as possible like one of die sensor array channels and has a value equal to die background capacitance of a typical sensor line, (i.e., wid no object proximate or present capacitance component). Capacitor 202 may be formed in several ways. Capacitor 202 may comprise an extra sensor line in a part of die sensor array, configured to approximate one of die active sensor lines but shielded from finger capacitance by a ground plane, etc. Alternately, capacitor 202 may be a capacitor formed in die integrated circuit or connected thereto and having a value selected to match that of a typical sensor line. In mis respect, the signal source comprising capacitor 202 and filter circuit 210 is somewhat like the circuitry for generating me ^ ma nd V m i n reference voltages, in that it mimics a typical sensor line. As another alternative, one of the actual sensor lines may be employed to set the bias voltage. The measured voltage on the two end-point sensor lines may be compared and the one having the lowest value may be selected on the theory that if a finger or other object is proximate to the sensor array, it will not be present at sensor lines located at die opposite edges of die array. According to anotiier aspect of me present invention, an "edge motion" feature may be implemented when die object position sensor of the present invention is used as a computer cursor control device in place of a mouse. A practical problem arises in the use of computer mice or other cursor control devices when an attempt is made to move an object over a large distance on a computer screen. This problem is encountered when a small mouse pad is used wid a computer mouse, or when an object position sensor of the kind described herein has a small touch sensor area. In touch sensor applications, this problem is especially acute during a "drag" gesture. If die user lifts die finger to begin a second stroke, die drag effect ends prematurely on die screen. The edge motion feature of the present invention helps to eliminate the need to use "rowing," or multiple strokes of the finger to move a large distance on me screen. A prior solution to the long-distance drag problem has been to provide an acceleration feature, i.e., a "ballistic" curve, where the gain varies as a function of finger speed, allowing die user to move long distances, albeit clumsily, using a repeated finger swishing motion. This technique can be used widi any variable-speed pointing device, for example, widi a mouse on a mouse pad of limited size. Typical mouse driver software includes an adjustable acceleration feature (sometimes under a misleading name like "mouse speed"). The system of the present invention can distinguish between d e commonly used "click", "double click" and "drag" mouse gestures. The single-click and double-click gestures are implemented in a mouse using the mouse to move die cursor to the object of interest on the computer screen and then "clicking" once or twice a mouse button connected to an electrical switch in the mouse body. A significant advantage of die present invention is that the need for mechanical switches is eliminated, and the single-click and double-click gestures are implemented in the present invention by moving the cursor to the object of interest and tiien tapping a finger tip once or twice on the selected object The present system can easily distinguish these gestures from the drag gesture in which the finger is placed on the touch sensor surface and dragged across die surface in a desired direction while maintaining contact with the surface. According to a presently preferred embodiment of the invention, the edge motion feature of the object position sensor is implemented by motion unit 18 of FIG. 1 and works by defining two zones in the sensing plane 10 containing the touch sensor array 22. As shown in FIG. 11, the sensing plane 10 is preferably divided into an inner zone 240 comprising most of the central portion of the surface of sensing plane 10 and an outer zone 242, typically comprising a thin marginal area at the periphery of the sensor array. The center of the sensing plane 10 may be described as the origin (X centef Y center ) m a cartesian coordinate system. Those of ordinary skill in the art will recognize however that the inner and outer zones could be of any shape. Thus in FIG. 11, inner zone 240 is defined by the upper dashed line Y 0 , right- hand dashed line X 0 , lower dashed line -Y 0 and left-hand dashed line -X Q . Outer zone 242 is the region between the outer edges of the sensing plane 10 defined by Y max , -Y mβΛ . X mκ and -X max and die outer borders of inner zone 240 defined by Y 0 , X Q , - Y Q , and -X Q . According to this aspect of the present invention, finger motions in the inner zone 240 are translated in the standard fashion into motion events to be sent to the host computer. As is well understood in the art, the standard way to communicate mouse motion to a host computer may also be employed in the present invention to communicate finger motion to a host computer. After the finger position is established as disclosed herein, die information communicated to die host computer is: ΔX = A(X cur - X old ) [Eq. 9] ΔY = A(Y cur - Y old ) [Eq. 10] where ΔX is die change in die X position of the finger, ΔY is the change in the Y position of the finger, X cur is die current X position of die finger and X old is die last reported X position of the finger, Y cur is the current Y position of die finger and Y 0 ι d is die last reported Y position of die finger, and A is a "gain factor" which is commonly encountered in mouse cursor control applications. Typically, the host computer takes (ΔX,ΔY) events and moves the cursor by the indicated amount in each axis, thus reconstructing the finger position on the screen as die successive ΔX and ΔY values are accumulated. So far, this is standard cursor control behavior where edge motion is not considered. According to the present invention, when the finger is reported as being in the outer zone 242, the edge motion feature of the present invention may be enabled. The determination of whether the finger is in the outer zone is a simple determination: [-Xo < X cur < X Q ] is FALSE, OR [-Y 0 < Y cur < Y r J is FALSE [Eq. 11 ] Referring now to FIG. 12, a circuit 244 for making a determination of whether a finger is in the outer zone 242 is shown in schematic diagram form. FIG. 12 illustrates a hardware embodiment for determining whedier a finger is in the outer zone 242, but those of ordinary skill in die art will readily recognize tiiat this determination could readily be made by performing one of a number of equivalent software routines. Such software routines are obvious and straightforward from the functions described herein. Circuit 244 includes digital comparators 246, 248, 250, and 252, which may be straightforwardly implemented by conventional logic. Comparator 246 puts out a true signal when the quantity X cur at one of its inputs is greater than the fixed quantity X Q presented to its otiier input. Comparator 248 puts out a true signal when the quantity X cur at one of its inputs is less than the fixed quantity -X 0 presented to its other input Comparator 250 puts out a true signal when the quantity Y cur at one of its inputs is greater than the fixed quantity Y 0 presented to its otiier input Comparator 252 puts out a true signal when the quantity Y cur at one of its inputs is less than the fixed quantity - Y 0 presented to its otiier input. The outputs of comparators 246, 248, 250, and 252 are ORed together by OR gate 254. As will be appreciated by those of ordinary skill in the art, the FingerOuter signal output of OR gate 254 is true only when the necessary conditions of Eq. 11 are satisfied. It is presentiy preferred tiiat die edge motion aspect of the present invention may be selectively enabled or disabled by a user. When the edge motion feature is enabled and the finger is reported as being in the outer zone as set forth above, a second component is added to die (ΔXΛY) events reported: ΔX =A (X cur - X old )+ S(X cur - X eβBtet ) [Eq. 12] ΔY = A(Y cur - Y old )+ S(Y cur - Y^) [Eq. 13] where X^-m er is the X coordinate of the center of the pad, Y center is the Y coordinate of the center of the pad, and S is a multiplicative factor for speed. S should be chosen such that the movement of the cursor is at a comfortable speed on the display screen. For example, if the finger is held a good distance to the right (so that X cur > XQ), then the cursor will tend to "glide" to the right at a constant speed set by multiplicative speed factor S in Eqs. 12 and 13. This factor can be adjusted to individual taste of a user. If the sensor array has different dimensions in X and Y, it is useful to set the multiplicative speed factor S parameters in the X and Y directions to differ by the same ratio as the pad dimensions, so that a finger held at the left or right edge of the sensor array will produce the same cursor speed as a finger held at the top or bottom edge. In the presendy preferred embodiment of die sensor array, tiiere are 24 X traces and 18 Y traces. Therefore, since X is 4/3 wider than Y (24 traces vs. 18 traces), the X multiplicative speed factor S x is set to be to be 3/4 as large as the multiplicative speed factor S γ . The glide speed of the cursor during edge motion is clearly a direct function of the distance of the finger from the center of the pad, and the glide direction is equal to the direction of the finger from the center. If the outer zone has the preferred "edge margin" shape as shown in FIG. 11, then the finger will always be roughly the same distance from the center whenever edge motion is activated (witiiin a factor of the square root of 2 = 1.41, assuming a square pad). Thus, the psychological effect is that edge motion involves a constant glide speed where the direction is set by the position around the sensor array edge. The square root of 2 variation may be canceled out by dividing the edge motion terms in equations (12 and 13) by a normalizing factor of the form: ^(X cur - Xcenter) 2 + (Y CU r " Ycenter) 2 [Eq. 14] but this is a computationally intensive step applied to fix a problem that is barely noticeable to die average user; tiius, it may be omitted. The edge motion feature of the present invention can be confusing if the user does not expect it Since edge motion is most useful in connection with the drag gesture, it is presendy preferred to arrange for it to occur only during a drag, i.e., only when die gesture logic is virtually "holding die mouse button down." The drag gesture and otiier gestures are implemented by gesture unit 20 of FIG. 1. At times when the edge-motion function is not desired, the outer zone 242 "goes away" (i.e., is ignored) and the inner zone 240 effectively expands to cover the entire sensing plane 10. It has been found that this is much less confusing in practice, probably because the user is more likely to be consciously aware of the cursor control device during a drag gesture than during simple cursor motions. Assuming the preferred zone boundary shape of FIG. 11, the following algorithm may be employed to implement the edge motion feature of the present invention: IF NOT (-X Q < X cur < X 0 AND -Y 0 < Y cur < Y 0 ) AND (optionally) a drag gesture in progress, THEN Let eX = S (X cur - X cen te r ) Let eY = Sγ(Y cur - Y ce n te r) ELSE Let eX = eY = 0. END IF Next die dX and dY motion terms are computed from the regular algoritiim: i.e., Let dX = A(X cur - X old ) Let dY = A(Y cur - Y old ) Finally, die resultant packet (ΔX = dX+eX, ΔY = dY+eY) is transmitted to die host computer. Those of ordinary skill in the art will recognize that a linear proportionality is described by die above equation. As used herein,"proportionality" means that the signal generated is a monotonic function. Those of ordinary skill in the art will recognize that other monotonic functions, including but not limited to inverse proportionality, and non-linear proportionality such as logaridimic or exponential functions, could be employed in the present invention without departing from the principles disclosed herein. A hardware implementation of this algorithm is illustrated in FIG. 13 in schematic diagram form. While circuit 256 is shown implemented in the X direction only, those of ordinary skill in die art will recognize that an identical circuit will also be employed in die Y direction. Such skilled persons will also immediately appreciate the complete equivalence of implementing the hardware solution of FIG. 13 as a software routine. Edge-motion circuit 256 includes a subtractor circuit 258 in which the previous value of X cur , stored in delay 260, is subtracted from the present value of X cur . The output o subtractor circuit 258 is presented to multiplier 262, which multiplies the result by the gain factor "A". The output of multiplier 262 is the term dX. The term X cur is also presented to subtractor circuit 264 in which the value of X center i s subtracted from the present value of X cur . The output of subtractor circuit 264 is presented to multiplier 266, which multiplies the result by the gain factor "S". A two-input AND gate 268 has its input terms the value FingerOuter from the circuit of FIG. 12 and the value MotionEnable which is a toggled on off enable signal for the edge motion feature of the present invention. If both FingerOuter and MotionEnable are true, switch 270 is configured to pass the output of multiplier 266 to adder circuit 272. If either FingerOuter or MotionEnable is false, then switch 270 is configured to pass die value zero to adder 272. The output of switch 270 is the eX term. The output of adder 272 is passed to die host computer as ΔX. The MotionEnable signal can be controlled by d e user, e.g., by a control panel. Alternatively, it may be controlled by d e gesture unit as will be more fully disclosed. In an alternate form, die dX term may be replaced by the eX term, and likewise for dY and eY, when the finger is in the "outer" zone, rather than adding die two terms in that zone. This results in a more "pure" edge motion which is harder for the user to guide. User tests have shown that the dX+eX form shown above feels better and is easier to use. Another alternative which is functional but has been found to be less desirable employs a somewhat wider outer zone. The glide speed is then varied in proportion to the distance of the finger into the outer zone rather than the distance from the center of the pad. Thus, as die finger enters die outer zone the glide speed starts at zero and increases to some reasonable limit as the finger reaches the edge of die pad. The result is a smoother transition between edge-motion and normal behavior. It is not difficult to modify d e above formulas to produce tiiis alternate behavior. This variant was tried by die inventors because die transition into edge-motion mode seemed too abrupt; tests showed that this abruptness is actually a boon in typical use. The smooth transition is harder to "feel", and thus winds up being more, not less, likely to catch the user by surprise. Those of ordinary skill in d e art will appreciate that a solution midway between die two described methods can also be employed to produce a less abrupt transition. An alternate solution to the long-distance drag problem has been to provide a "locking" drag, but this can be confusing since the lock state becomes a hidden mode, a well- known undesirable item in the study of user interfaces. The edge motion feature of the present invention is used advantageously with one or more finger gestures which may be performed by a user on the sensor array and are recognized by the system. Of particular interest are the basic tap and drag gestures. The tap gesture is analogous to the clicking of the mouse button on a conventional mouse, and the concept of dragging objects is familiar to all mouse users. The drag gesture consists of tapping once, followed by bringing the finger back down to the touch surface and moving it in the desired fashion. The host computer sees the virtual mouse button being held down during this entire gesture. Referring back to FIG. 1, gesture unit 20 detects the tap and drag gestures as follows. There are four state variables used to recognize the tap and drag gestures: TapState, NONE if there is no gesture in progress, TAP if a tap gesture is in progress, or DRAG if a drag gesture is in process; TapOkay, TRUE if high enough Z (finger pressure) has been seen to qualify as a tap; DownTime, the time when the finger last touched down on the sensor pad; and DownPos, the starting (X,Y) position of the current tap. The parameters utilized are: Tap Time is the maximum duration of a tap gesture; DragTime is the maximum duration of a tap-and-drag gesture; TapRadius is the maximum distance moved during a tap gesture; Ztap is die minimum pressure (Z) for a tap gesture; and Zthresh is die minimum pressure (Z) to detect a finger. A process according to die present invention for recognizing tap and drag gestures is illustrated in die flow diagrams of FIGS. 14a- 14c. Processing begins at step 280 for every new set of (X,Y,Z) data to arrive from die aridimetic unit 16 of FIG. 1. In a current embodiment of die system, such data arrive 40 times per second. In step 282, the Z (pressure) value is compared against the Z threshold Zthresh to determine whetiier a finger is present ("down") or not ("up"). Instead of a simple tiireshold comparison, two thresholds may be used to provide hysteresis as is well-known in d e art In step 284, the finger is known to be down. The previous Z is checked to see whedier the finger was previously down or is just now touching down on die pad. In step 286, a finger-down transition has been detected. If a tap gesture has recently been reported (Tapstate = TAP), tiien, in step 288 this pending tap gesture is converted into a potential drag gesture. Step 290 records die position and the time at which the finger touched down. Step 292 initializes the TapOkay flag, which records whedier or not the Z (pressure) information ever exceeds the Ztap threshold during the current tap. In step 294, the finger is known to be up. The previous Z is checked to see whether the finger was previously up or is just now being lifted from the pad. In step 296, a finger-up transition has been detected. Various tests are made of the most recent finger-down period to see if it qualifies as a tap. To qualify, the finger-down period must have short duration (CurTime minus DownTime must be less than TapTime), little or no motion (die distance from CurPos to DownPos must be less than TapRadius) , and sufficient peak finger pressure (TapOkay must be TRUE), in order to qualify. During die finger-down period which has qualified as a tap, a small amount of motion may have occurred due to natural vibration of the finger and otiier variations. This motion will have already been transmitted to the host, causing the cursor to drift slightiy away from the intended target of d e tap gesture. In step 298, this spurious motion is replayed to die host in reverse in order to cancel it out. The motion may be sent in a single lump-sum packet or replayed packet by packet. If die host computer is not expected to apply extreme non-linear gain ("ballistics") to the received packets, a simple lump-sum transmission will suffice. Steps 300 and 302 give special consideration to multiple taps. If a tap gesture is recognized, and a previous tap gesture (at this point a potential drag gesture) is pending, then one or more packets reporting a release of the virtual mouse button are sent to the host This prevents the virtual button presses resulting from the two taps from running together into a single long virtual button press. The special button-release packet may be accomplished by sending an extra packet to the host or by setting a flag to suppress the virtual mouse button for the next subsequent packet. Step 304 records that a tap gesture is now in progress. In step 306, the finger is still away from the pad. If die finger has been away from die pad long enough so that the current tap gesmre (if any) no longer qualifies to be extended into a drag (i.e., Curtime minus DownTime exceeds DragTime), or if a drag gesture is pending (TapState is DRAG), then the current tap or drag is terminated in step 308 (causing die virtual mouse button to be released). Steps 310 and 312 compare Z against Ztap. If Z is ever greater than Ztap during a given finger-down period, die TapOkay flag is set. Steps 314, 316, and 318 cause die virtual mouse button to be pressed during a tap or drag gesture, and released at all other times. In a current embodiment of an actual system according to the present invention, TapTime is a fraction of a second and may be adjusted by the user, DragTime is approximately equal to TapTime, and TapRadius is approximately 5-10% of the pad width. The "MotionEnable" signal into AND gate 268 of FIG. 13 is obtained by either MotionEnable = (TapState = DRAG) or MotionEnable = (TapState = DRAG) OR (TapState = TAP). While the foregoing disclosure refers to two particular gestures, those of ordinary skill in the art will recognize that other gestures may be employed in the system of die present invention. The increased sensitivity of the touch sensor system of the present invention allows for a lighter input finger touch which makes it easy for human use. Increased sensitivity also makes it easier to use other input objects, like pen styli, etc. Additionally this sensitivity allows for a trade-off against a thicker protective layer, or different materials, which both allow for lower manufacturing costs. Greater noise rejection allows for greater flexibility in use and reduced sensitivity to spurious noise problems. Two techniques are employed which allow derivation of die most noise-rejection benefit Due to the drive and sense techniques employed in the present invention, the data acquisition rate has been increased by about a factor of 30 over die prior art This offers several obvious side effects. First, for die same level of signal processing, die circuitry can be turned off most of die time and reduce power consumption by roughly a factor of 30 in die analog section of the design. Second, since more data is available, more signal processing, such as filtering and gesture recognition, can be performed. The sensor electronic circuit employed in the present invention is very robust and calibrates out process and systematic errors. It will process the capacitive information from the sensor and provide digital information to an external device, for example, a microprocessor. Because of the unique physical features of the present invention, there are several ergonomically interesting applications that were not previously possible. Presently a mouse or trackball is not physically convenient to use on portable computers. The present invention provides a very convenient and easy-to-use cursor position solution that replaces those devices. In mouse-type applications, the sensor of the present invention may be placed in a convenient location, e.g., below the "space bar" key in a portable computer. When placed in this location, the thumb of the user may be used as the position pointer on the sensor to control the cursor position on the computer screen. The cursor may then be moved without the need for the user's fingers to leave the keyboard. Ergonomically, this is similar to the concept of the Macintosh Power Book with it's trackball, however the present invention provides a significant advantage in size over die trackball. Extensions of this basic idea are possible in that two sensors could be placed below die "space bar" key for even more feature control. The computer display widi it's cursor feedback is one small example of a very general area of application where a display could be a field of lights or LED's, an LCD display, or a CRT. Examples include touch controls on laboratory equipment where present equipment uses a knob button/touch screen combination. Because of the articulating ability of is interface, one or more of those inputs could be combined into one of die inputs described widi respect to die present invention. Consumer Electronic Equipment (stereos, graphic equalizers, mixers) applications often utilize significant front panel surface area for slide potentiometers because variable control is needed. The present invention can provide such control in one small touch pad location. As Electronic Home Systems become more common, denser and more powerful human interface is needed. The sensor technology of the present invention permits a very dense control panel. Hand-held TV/VCR/Stereo controls could be ergonomically formed and allow for more powerful features if this sensor technology is used. The sensor of the present invention can be conformed to any surface and can be made to detect multiple touching points, making possible a more powerful joystick. The unique pressure detection ability of die sensor technology of die present invention is also key to tiiis application. Computer games, "remote" controls (hobby electronics, planes) , and machine tool controls are a few examples of applications which would benefit from the sensor technology of d e present invention. Musical keyboards (synthesizers, electric pianos) require velocity sensitive keys which can be provided by die pressure sensing ability of this sensor. There are also pitch bending controls, and other slide switches tiiat could be replaced widi this technology. An even more unique application comprises a musical instrument that creates notes as a function of the position and pressure of the hands and fingers in a very articulate 3-d interface. The sensor technology of the present invention can best detect any conducting material pressing against it. By adding a compressible insulating layer covered by a layer of conductive material on top of the sensor the sensor of the present invention may also indirectly detect pressure from any object being handled, regardless of its electrical conductivity. Because of die amount of information available from this sensor it will serve very well as an input device to virtual reality machines. It is easy to envision a construction mat allows position-monitoring in three dimensions and some degree of response (pressure) to actions. While embodiments and applications of this invention have been shown and described, it would be apparent to those skilled in die art tiiat many more modifications than mentioned above are possible without departing from the inventive concepts herein. The invention, therefore, is not to be restricted except in the spirit of die appended claims.

Claims: Claims We claim: 1. A method for providing an electrical signal for moving a cursor on a display screen associated with a computer in response to electrical signals representing of the position of an object in a two dimensional sensing plane, including the steps of: providing a sensing plane including a matrix of conductors arranged as a plurality of rows and columns of spaced apart row conductive lines and column conductive lines, said sensing plane characterized by an inherent capacitance on the various ones of said row conductive lines and column conductive lines, said capacitance varying widi the proximity of an object to said row and column conductors; sensing the capacitance on at least selected ones of said row and column conductors; generating present-position signals from the sensed capacitance representing the present position of said object in both X and Y directions on said sensing plane; sensing whether said object is in a peripheral region of said sensing plane; generating first relative position X and Y signals representing the difference between said present position of said object in both X and Y directions and a previous position of said object in both X and Y directions; sending said first relative position X and Y signals to said computer if said object is not in said peripheral region of said sensing plane; generating second relative position X and Y signals if said object is in a peripheral region of said sensing plane; sending said second relative position X and Y signals to said computer so long as said object is in said peripheral region of said sensing plane. 2. The method of claim 1 wherein said present-position signals, said first relative position X and Y signals, and said second relative position X and Y signals are digital signals. 3. A method for providing an electrical signal for moving a cursor on a display screen associated widi a computer in response to electrical signals representing of the position of an object in a two dimensional sensing plane, including the steps of: providing a sensing plane including a matrix of conductors arranged as a plurality of rows and columns of spaced apart row conductive lines and column conductive lines, said sensing plane characterized by an inherent capacitance on the various ones of said row conductive lines and column conductive lines, said capacitance varying with the proximity of an object to said row and column conductors; sensing the capacitance on at least selected ones of said row and column conductors; generating present-position signals from the sensed capacitance representing die present position of said object in both X and Y directions on said sensing plane; sensing whether said object is in a peripheral region of said sensing plane; generating first relative position X and Y signals representing the difference between said present position of said object in both X and Y directions and a previous position of said object in both X and Y directions; sending said first relative position X and Y signals to said computer if said object is not in said peripheral region of said sensing plane; generating second relative position X and Y signals if said object is in a peripheral region of said sensing plane, said second relative position X signal comprising said first X signal incremented by an amount proportional to the difference in said X direction between said present position in said X direction of said object and a fixed X position on said sensing plane, said second relative position Y signal comprising said first Y signal incremented by an amount proportional to the difference in said Y direction between said present position in said Y direction of said object and a fixed Y position on said sensing plane; sending said second relative position digital X and Y signals to said computer so long as said object is in said peripheral region of said sensing plane. 4. The metiiod of claim 3 wherein said present-position signals, said first relative position X and Y signals, and said second relative position X and Y signals are digital signals. 5. The method of claim 3 wherein: said amount proportional to the difference in said X direction between said present position in said X direction of said object and a fixed X position on said sensing plane is m times the difference in said X direction between said present position in said X direction of said object and a center X position on said sensing plane; and said amount proportional to the difference in said Y direction between said present position in said Y direction of said object and a fixed Y position on said sensing plane is n times the difference in said Y direction between said present position in said X direction of said object and a center Y position on said sensing plane; wherein m and n are numbers chosen to impart a desired speed to die motion of said cursor on said display. 6. The method of claim 5 wherein the ratio of m to n is equal to the ratio of the width of said sensing plane to die height of said sensing plane. 7. A method for providing an electrical signal for moving a cursor on a display screen associated widi a computer in response to electrical signals representing of the position of an object in a two dimensional sensing plane, including die steps of: providing a sensing plane including a matrix of conductors arranged as a plurality of rows and columns of spaced apart row conductive lines and column conductive lines, said sensing plane characterized by an inherent capacitance on die various ones of said row conductive lines and column conductive lines, said capacitance varying with die proximity of an object to said row and column conductors; simultaneously developing a first set of signals proportional to the value of said capacitance for each of said row conductive lines when no object is located proximate to said sensing plane; simultaneously developing a second set of signals proportional to the value of said capacitance for each column conductive lines when no object is located proximate to said sensing plane; simultaneously developing a third set of signals proportional to the value of said capacitance for each of said row conductive lines when an object is located proximate to said sensing plane; simultaneously developing a fourth set of signals proportional to the value of said capacitance for each column conductive lines when said object is located proximate to said sensing plane; computing a first weighted average of the difference between said first set of signals and said third set of signals to generate a present-position signal in the X direction of said sensing plane; and computing a second weighted average of the difference between said second set of signals and said fourth set of signals generate a present-position signal in the Y direction of said sensing plane; generating first relative position X and Y signals representing the difference between said present-position signals in both X and Y directions and a previous set of present-position signals in both X and Y directions; sending said first relative position X and Y signals to said computer if said object is not in said peripheral region of said sensing plane; generating second relative position X and Y signals if said object is in a peripheral region of said sensing plane; sending said second relative position X and Y signals to said computer so long as said object is in said peripheral region of said sensing plane. 8. The method of claim 7, wherein the steps of simultaneously developing said first, second, third, and fourth sets of signals includes the steps of: placing a first known voltage on said capacitances; discharging said capacitances for a fixed time at a fixed current; measuring and storing a first set of resultant voltages across said capacitances; placing a second known voltage on said capacitances; charging said capacitances for said fixed time at said fixed current; measuring and storing a second set of resultant voltages across said capacitances; and averaging corresponding ones of said first and second sets of resultant voltages. 9. The method of claim 7 wherein the steps of computing said first and second weighted averages comprises the steps of: computing a sum and a weighted sum of said first set of signals; computing a sum and a weighted sum of said second set of signals; computing a sum and a weighted sum of said third set of signals; computing a sum and a weighted sum of said fourth set of signals; computing a row numerator by subtracting said weighted sum of said first set of signals from said weighted sum of said third set of signals; computing a row denominator by subtracting said sum of said second set of signals from said sum of said fourth set of signals; dividing said row numerator by said row denominator to derive a row position signal representing the position of said object in a row dimension; computing a column numerator by subtracting said weighted sum of said second set of signals from said weighted sum of said second set of signals; computing a column denominator by subtracting said sum of said second set of signals from said sum of said second set of signals; and dividing said column numerator by said column denominator to derive a column position signal representing the position of said object in a column dimension. 10. The method of claim 9 including the further steps of: storing said sum and said weighted sum of said first and third set of signals as a stored sum and a stored weighted sum of said second and fourth sets of signals; and using said stored sum and said stored weighted sum in computing subsequent ones of said row numerators and denominators and said column numerators and denominators using said stored sum and a stored weighted sum for providing an electrical signal representative of a subsequent position of said object in said two dimensional plane.

Inventors: ALLEN TIMOTHY P, FAGGIN FEDERICO, GILLESPIE DAVID, MILLER ROBERT J, ALLEN, TIMOTHY, P., FAGGIN, FEDERICO, GILLESPIE, DAVID, MILLER, ROBERT, J.
Applicants: ALLEN TIMOTHY P, FAGGIN FEDERICO, GILLESPIE DAVID, MILLER ROBERT J, SYNAPTICS INC, ALLEN, TIMOTHY, P., FAGGIN, FEDERICO, GILLESPIE, DAVID, MILLER, ROBERT, J., SYNAPTICS, INC.
Application year: 1995
Main classification: G06F3
IPC codes: G06F3