Driving circuit for light emitting diode

Abstract: A circuit having a differential pair of bipolar transistors (1,2) and a constant current source (3). A series circuit consisting of a resistor (109) and a capacitor (110) is connected between the collector of one of the resistors and a steady voltage source either that of the power supply or of the substrate. The differential pair, constant current source, and series circuit are formed on an integrated circuit in a semiconductor chip.

Description: The present invention relates to a driving circuit for driving a semiconductor light emitting element such as a light emitting diode (LED) or a semiconductor laser. Such driving circuits may be used for driving LED displays, LED indicators, or LED printer heads, as laser source driving circuits for optical disk apparatuses, exposure apparatuses for IC fabrication, or laser beam printers, or as driving circuits for light emitting elements for optical communications. Driving circuits for semiconductor light emitting elements represented by light emitting diodes (LEDs) and semiconductor lasers are classified into two categories: a cathode common circuit and an anode common circuit. In the cathode common circuit, the cathode of a light emitting element is connected to a low-potential reference voltage source, and its anode is connected to a constant current source. In the anode common circuit, the anode of a light emitting element is connected to a high-potential reference voltage source, and its cathode is connected to a switching circuit and a constant current source. One example of the former circuit is disclosed in Japanese Laid-Open Patent Application No. 2-296382. The latter circuit is expected as a future mainstream circuit since its switching speed is higher than that of the former. Fig. 1 is a circuit diagram showing an example of a conventional semiconductor light emitting element driving circuit. Referring to Fig. 1, this driving circuit consists of differentially connected bipolar transistors 1 and 2; a bipolar transistor 3 which operates with a constant current; a semiconductor light emitting element 4; a resistor 5; an inductance 6 of the line which connects the bipolar transistors for driving the semiconductor light emitting element and the semiconductor light emitting element; an inductance 7 of the line which connects the semiconductor light emitting element and a power supply; a junction capacitance 8 of the semiconductor light emitting element; collector-emitter capacitances 9 and 10 of the bipolar transistors 1 and 2, respectively; and a base-collector capacitance 11 of the bipolar transistor 3. As in Fig. 1, in this semiconductor light emitting element driving circuit constituted by a conventional semiconductor integrated circuit, the emitters of the differentially connected bipolar transistors 1 and 2 are connected to the collector of the bipolar transistor 3 which has the same conductivity type and operates as a constant current source. Unfortunately, in an arrangement such as this prior art shown in Fig. 1 the current waveform for driving a semiconductor light emitting element is in some instances significantly distorted. Fig. 2 is a view for explaining the mechanism in which the driving current for the semiconductor light emitting element is distorted in the configuration illustrated in Fig. 1. This mechanism will be described below with reference to Fig. 2. The semiconductor light emitting element 4 needs to be switched at a high speed with a constant current of about a few milliamperes to 100 milliamperes. Assume that the value of this constant current is I and the values of the inductances 6 and 7 are L 1 and L 2 , respectively. Assuming the time during which the transistor 1 is activated from a cut-off state and the current flowing through the semiconductor light emitting element 4 reaches the constant current value I from substantially zero is t , counterelectromotive voltages V 1 and V 2 produced in the inductances 6 and 7, respectively, by this current change are V 1 = L 1 ·I/t, V 2 = L 2 ·I/t V 1 is transmitted to the base of the transistor 3 through the collector-emitter capacitance 9 of the transistor 1 and the base-collector capacitance 11 of the transistor 3. Likewise, V 2 is transmitted to the base of the transistor 3 through the junction capacitance 8 of the semiconductor light emitting element 4 and the capacitances 9 and 11. The mutual conductance, g m , of the bipolar transistors is expressed as follows: g m = ∂I C /∂V BE = qI c /KT That is, a change in the potential applied to the base produces as a large change in the collector current. In this conventional arrangement, therefore, a potential variation occurring in the line inductance is transmitted to the base of the transistor 3 which operates with a constant current, thus distorting the collector current, i.e., the driving current.waveform of the semiconductor light emitting element. This distortion in the waveform is represented by overshoot or ringing. If overshoot takes place, a semiconductor light emitting element deteriorates, and this consequently reduces the life of a product using this semiconductor light emitting element. Examples of techniques for preventing this include circuit configurations shown in Figs. 3 and 4. In Fig. 3, a snubber circuit consisting of a resistor 12 and a capacitor 13 is additionally provided externally to the package incorporating a semiconductor light emitting element driving circuit. In Fig. 4, the emitters that are differentially connected and the base and collector of the transistor which operates with constant current are extracted to terminals, and a coil 14 or a capacitor 24 is inserted to stabilise the constant current. In Figs. 3 and 4, connecting terminals are denoted by 19 to 23. The technique illustrated in Fig. 3, however, delays the rise time due to the resistor and the capacitor for removing overshoot, leading to a decrease in the driving frequency. In the technique illustrated in Fig. 4, the number of package pins and the number of parts are increased. In a conventional circuit for switching the current of a semiconductor light emitting element at a high frequency, a series circuit of a resistor 109 and a capacitor 110 as discrete circuit components, Fig. 5, are provided outside a package 116 which incorporates an integrated circuit 117, as illustrated in Fig. 6. Fig. 5 is an equivalent circuit diagram of this prior art which includes parasitic elements (parasitic inductance components). Fig. 6 shows the packaged state of the circuit shown in Fig. 5. Referring to Figs. 5 and 6, this circuit includes differentially connected bipolar transistors 1 and 2 formed in the integrated circuit 117; a constant current source 30; a semiconductor light emitting element 4; a resistor 5; a parasitic inductance 106 of the bonding wire for -- connecting the collectors of the bipolar transistors for driving the semiconductor light emitting element to the pin of the package 116; a parasitic inductance 107 of the pin; a parasitic inductance 108 of the line from the pin to the semiconductor light emitting element 4; the resistor 109; the capacitor 110; a parasitic inductance 112 resulting from the series connection of the resistor 109 and the capacitor 110; a power supply 15; complementary pulse signals 114 and 115 for driving the transistors 1 and 2; and the integrated circuit 117 including the transistors 1 and 2. A semiconductor light emitting element needs to be switched at a high speed with a constant current of about a few milliamperes to 100 milliamperes. Assume that the value of this constant current is I and the values of the parasitic inductances 106, 107, 108, and 112 are L 1 , L 2 , L 3 , and L 4 , respectively. Fig. 7 is an approximate circuit in which the transistor 1, Fig. 5, is represented by a switch 118, and L 1 = L 2 = 0 [H]. In Fig. 7, when the switch 118 is closed at time t = 0, the following equations are established at a point A: V A = 1/C·∫ i 2 dt + L 4 ·di 1 /dt + Ri 2 V A = V CC - L 3 ·di 1 /dt i 1 = i 2 + I where V A is the potential at the point A and V CC is the power supply voltage. By substituting Equation 3 into Equation 1 and differentiating, Equation 4 below is obtained: dV A /dt = 1/C(i 1 - I) + L 4 ·d 2 i 1 /dt 2 + R·di 1 /dt In addition, Equation 5 below is obtained by differentiating Equation 2: dV A /dt = -L 3 ·d 2 i 1 /dt 2 From Equations 4 and 5, Equation 6 below results: (L 3 + L 4 )·d 2 i 1 /dt 2 + R·di 1 /dt + 1/C(i 1 - I) = 0 Characteristic equation (D) of Equation 6 is represented by (D) = (L 3 + L 4 )D 2 + R·D + 1/C Assuming the roots of (D) are λ 1 and λ 2 , a transient solution is as follows: i 1 = (λ 1 t)th power of K 1 ·e + (λ 2 t)th power of K 2 ·e (K 1 and K 2 are constants) In a steady state after the switch 118 is closed, i 1 = I. Therefore, a general solution is given by Equation 9 below: i 1 = I + (λ 1 t)th power of K 1 ·e + (λ 2 t)th power of K 2 ·e Solving Equation 7 yields Equation 10 below: D = -R/2(L 3 + L 4 ) ± j√(ω 0 2 - τ -2 ) = -τ -1 ± jω f = λ 1 , λ 2 where ω o = 1/√(L 3 + L 4 )C), τ = 2(L 3 + L 4 )/R, ω f = √(ω o 2 - τ -2 ) In Equation 10, since the value inside √ can be either positive or negative, the following three cases are possible: (1) ω 0 > τ -2 (2) ω 0 < τ -2 (3) ω 0 = τ -2 In case (1), both λ 1 and λ 2 are complex numbers, so i 1 becomes oscillatory. Differentiating Equation 9 yields Equation 11 below: di 1 /dt = (λ 1 t)th power of K 1 λ 1 ·e + (λ 2 t)th power of K 2 λ 2 ·e Assuming in Equations 9 and 11 that, for t = 0, i 1 = 0, t = ∞, and i 1 = I, 0 = I + K 1 + K 2 and 0 = K 1 λ 1 + K 2 λ 2 Therefore, K 1 = λ 2 I/(λ 1 - λ 2 ) and K 2 = -λ 1 I/(λ 1 - λ 2 ) Consequently, Equation 12 below is obtained: i 1 = I[1 - {-(t/τ)th power of ω 0 /ω f ·e}·sin(ω f t + )]  = tan -1 ω f τ Equation 12 indicates that the oscillation of the frequency ω f logarithmically decreases according to the time constant τ. In case (2), both λ 1 and λ 2 are negative real numbers, so an unoscillating state results. λ 1 , λ 2 = -τ -1 ± ω s where ω s = jω f , ω f = √[{R/2(L 3 + L 4 )} 2 - {1/√((L 3 + L 4 )C)} 2 ] The result is Equation 13 below: i 1 = I[1 - {-(t/τ)th power of ω 0 /ω s ·e}·sin(ω s t + )]  = tan -1 ω s τ Consequently, i 1 changes logarithmically. In case (3), λ 1 = λ 2 = -τ -1 , which means a critical point. For ω f = 0, ω 0 = τ -1 i 1 = I[1 - {-(t/τ)th power of ω 0 /ω s ·e}·sin(ω f t + )] For ω f → 0, therefore, i 1 = I{1 + (-t/τ)th power of (1 + t/τ)·e}. In cases (1), (2), and (3), the waveforms of the current flowing through the semiconductor light emitting element are as indicated by (a), (b), and (c), respectively, in Figs. 8A to 8C. The waveform shown in Fig. 8A degrades the life of the semiconductor light emitting element because of occurrence of overshoot, leading to a short product life. The waveform shown in Fig. 8B increases the rise time to make high-speed switching impossible. The most desired waveform is as illustrated in Fig. 8C. In this case, Equation 14 below is obtained from Equation 11: 1/√((L 3 + L 4 )C ) = R/2(L 3 + L 4 ) Consequently, the resistance of the resistor 109 and the capacitance of the capacitor 110 with which the rise time can be minimized can be determined from the values of the parasitic inductances L 3 and L 4 . Here, L 1 = L 2 = 0 [H] assumed in the above quantitative consideration is important. Since it is assumed that L 1 = L2 = 0 [H], it is possible to determine the resistance of the resistor 109 and the capacitance of the capacitor 110 with which the rise time is minimized and no overshoot takes place. However, since conventional circuits make use of a resistor and a capacitor as discrete parts external to the package, the parasitic inductance 106 (= L 1 ) of the bonding wire and the parasitic inductance 107 (= L 2 ) of the pin of the package exist. Consequently, L 1 = L 2 = 0 [H] cannot be established. This makes it difficult to prevent overshoot. To prevent overshoot, the capacitance of the capacitor 110 must be increased. Unfortunately, increasing the capacitance of the capacitor 110 lowers the operating speed. The present invention is intended to provide a semiconductor light emitting element driving circuit capable of driving a semiconductor light emitting element with a stable driving current by eliminating distortion of the waveform of the driving current for the semiconductor light emitting element, thereby preventing deterioration of the semiconductor light emitting element and increasing the life of a product. The present invention is also intended to provide a semiconductor light emitting element driving circuit capable of preventing a decrease in the driving frequency or increases in the number of package pins and the number of parts. An embodiment of the present invention, as defined in claim 1 appended hereto, provides a circuit in which a snubber circuit consisting of a resistor and a capacitor in series is formed in an integrated circuit, in which differentially connected bipolar transistors are formed, and is connected between the collectors of the bipolar transistors and a power supply or a substrate potential. With this arrangement, the circuit can effectively suppress overshoot and prevent a decrease in the operating speed. In the accompanying drawings: Fig. 1 is a circuit diagram showing a conventional semiconductor light emitting element driving circuit; Fig. 2 is a view showing voltage waveforms and current waveforms at individual nodes when the circuit in Fig. 1 is in operation; Fig. 3 is a circuit diagram showing another driving circuit; Fig. 4 is a circuit diagram showing still another driving circuit; Fig. 5 is a circuit diagram showing a conventional semiconductor light emitting element driving circuit; Fig. 6 is a view showing the packaged state of the circuit in Fig. 5; Fig. 7 is a simplified circuit diagram of Fig. 5; Figs. 8A to 8C are views showing driving current waveforms; Fig. 9 is a circuit diagram showing a semiconductor light emitting element driving circuit according to a first background example; Fig. 10 is a view for explaining voltage waveforms and current waveforms at individual nodes when the circuit in Fig. 9 is in operation; Figs. 11A to 11C are views showing semiconductor light emitting element driving current waveforms and constant current NMOS gate voltage waveforms; Fig. 12 is a circuit diagram showing a driving circuit according to a second background example; Fig. 13 is a circuit diagram showing a driving circuit according to an embodiment of the present invention; Fig. 14 is a sectional view showing a portion of a driving circuit chip according to the embodiment of the present invention shown in Fig. 13; Fig. 15 is a sectional view showing another portion of the driving circuit chip according to the embodiment of the present invention shown in Fig. 13; and Figs. 16A and 16B are schematic views showing the arrangements of systems using the driving circuit of the present invention. In the present invention a switching circuit consisting of a pair of differentially connected bipolar transistors and a snubber circuit consisting of a resistor and a capacitor in series are integrated into a one-chip IC. This circuit is more preferably a driving circuit in which a driving means having a switching circuit consisting of a pair of differentially connected bipolar transistors and a constant current circuit constituted by a MOS transistor is connected to the cathode of a light emitting element. In this circuit configuration, the snubber circuit of the resistor and the capacitor in series is connected to the cathode of the light emitting element, and the driving circuit and the snubber circuit are integrated into one chip. These driving circuits can be realized as a one-chip IC by using well-known IC process techniques. It is also possible to fabricate the circuits by using a compound semiconductor such as gallium arsenide or indium phosphorus. When a driving circuit is to be fabricated by using the same compound semiconductor as a light emitting element, it is readily possible to integrate the two into one chip. Fig. 9 is a circuit diagram showing a semiconductor light emitting element driving circuit according to a first background example. Referring to Fig. 9, npn transistors 1 and 2 are differentially connected. An NMOS transistor 31 has a drain connected to the common emitter of the npn transistors 1 and 2 and operates with a constant current. A semiconductor light emitting element 4 has a cathode connected to the collector of the npn transistor 1 via a parasitic inductance 6 of a line. A resistor 5 as a load is connected to the collector of the npn transistor 2. A parasitic inductance 7 is due to the line between the anode of the semiconductor light emitting element 4 and a power supply 15 as a high-potential reference voltage source. The semiconductor light emitting element 4 has a junction capacitance C j , 8. The npn transistors 1 and 2 have emitter-collector capacitances 9 and 10, respectively. The NMOS transistor 3 which operates as a constant current source has a drain-gate capacitance 11. One end of a resistor 41 is connected to the gate of the NMOS transistor 3 and to one end of a capacitor 131 The other end of the resistor 41 is connected to the gate and drain of an NMOS transistor 53 for applying a bias potential. The other end of the capacitor 131 is connected to a ground potential 16 as a low-potential reference voltage source. Inverters 17 and 18 supply complementary switching signals to the bases of the differentially connected npn transistors. A constant current source 51 supplies a constant current to the NMOS transistor 53. Fig. 10 is a view showing voltage waveforms and current waveforms at individual nodes in the first background example. The operation of this example will be described below with reference to Fig. 10. The inverter 17 applies a High-level signal to the base of the npn transistor 1, and the inverter 18 applies a Low-level signal to the base of the npn transistor 2. Consequently, the npn transistor 1 is activated, and the npn transistor 2 is cut off. The collector current of the npn transistor 1, i.e., the driving current of the semiconductor light emitting element changes from 0 to a drain current I of the NMOS transistor 31 which operates with a constant current, in a certain time t. As indicated by (a) and (b) in Fig. 10, the current change I in the time t generates counterelectromotive voltage pulses L 1 ·I/t and L 2 ·I/t due to the parasitic inductances 6 (= L 1 ) and 7 (= L 2 ), respectively. These counterelectromotive voltage pulses are transmitted to the gate of the NMOS transistor 31 through the junction capacitance 8 of the semiconductor light emitting element, the emitter-collector capacitance 9 of the npn transistor 1, and the gate-drain capacitance 11 of the NMOS transistor 3. Potential variations at the emitters of the npn transistors 1 and 2 are illustrated in (e) of Fig. 10, i.e., indicated by the solid-line waveform which is generated by the complementary driving signals to the bases and the dotted-line waveform which is synthesized with the counterelectromotive voltage pulses, respectively. Consequently, the dotted-line potential waveform shown in (f) of Fig. 10 is generated at the gate of the NMOS transistor. The mutual inductance, g m , of the NMOS transistor is represented by g m = √(I D ·µ n C ox W/L) where I D : the drain current µ n : the mobility of electrons C ox : the gate capacitance W : the channel width L : the channel length The above equation demonstrates that the mutual conductance of a MOS transistor is far smaller than that of a bipolar transistor. This indicates that the influence of a potential variation at the gate is small as a change in the drain current. Furthermore, in this embodiment the resistor 41 and the capacitor 131 are connected to the gate of the NMOS transistor 3. Consequently, the behavior of the gate potential is controlled by time constant τ = C G -R G . That is, the gate potential responds to high-speed pulses by the time constant τ. In Figs. 11A to 11C indicate the driving current waveforms of the semiconductor light emitting element and gate voltage waveforms obtained while changing the time constant obtained by the resistor 12 and the capacitor 13 connected to the gate of the NMOS transistor 3 which operates with a constant current. Figs. 11A to 11C show that when the time constant is made most suitable, a current waveform free from overshoot and ringing can be obtained. It is preferable to realise the MOS transistor 31 by connecting MOS transistors in parallel, which are individually formed in a plurality of active regions formed on an Si substrate and isolated by a field insulating film made from SiO 2 as a dielectric substance. Fig. 12 is a circuit diagram showing a semiconductor light emitting element driving circuit according to a second background example. In this example, a plurality of NMOS transistors 31,32,...,3n are connected in parallel, between a common current line 57 and the ground line 16, and a respective resistor 41,42,...4n and a respective capacitor 131,132,...,13n are connected to the gate of each MOS transistor. In this arrangement the other end of each resistor 41,42,...4n is connected to a common voltage line 55, and the other end of- each capacitor 131,132,...,13n is connected to the ground line 16. Also as shown in fig. 12, a current source 51 is connected between voltage lines 15 and 55, and an NMOS transistor 53 is connected between voltage line 55 and the ground line 16, with its gate also connected to voltage line 55. This is so because, if the W/L of an NMOS transistor is increased to obtain a necessary driving current, the behaviour of the gate must be considered as a distributed constant circuit, i.e. it is not possible to handle the transistor as a lumped constant, and this makes an optimum design for the circuit difficult. As discussed above, a MOS transistor is used as a constant current source connected to the emitters of differentially connected bipolar transistors. Additionally, a time constant circuit consisting of a resistor and a capacitor is connected to the gate of this MOS transistor. Consequently, it is possible to obtain a current pulse waveform free from overshoot and ringing. Since a semiconductor light emitting element is driven with a stable driving current, the element does not deteriorate. This effectively increases the life of the product. It is also possible to solve the problem of a decrease in the driving frequency which is due to conventional countermeasures or the problem of an increase in the number of package pins and the number of parts. First Embodiment A semiconductor light emitting element driving circuit according to a first embodiment of the present invention will be described below. Fig. 13 is a circuit diagram showing the semiconductor light emitting element driving circuit. Referring to Fig. 13, this driving circuit comprises differentially connected bipolar transistors 1 and 2 formed in an integrated circuit (not shown), a constant current source 30, a semiconductor light emitting element 4, a resistor 5, parasitic inductances 106, 107, and 108, a resistor 109, and a capacitor 110. The parasitic inductance 106 is due to the bonding wire connecting the collectors of the bipolar transistors for driving the semiconductor light emitting element to the pin of a package 116. The parasitic inductance 107 is due to the pin, and the parasitic resistance 108 is derived from the line from the pin to the semiconductor light emitting element 4. The resistor 109 is formed in the integrated circuit in which the bipolar transistors are formed. The capacitor 110 is also formed in the same integrated circuit. The resistor 109 and the capacitor 110 are connected in series in the integrated circuit. One terminal of this series circuit is connected to the collectors of the bipolar transistors for driving the semiconductor light emitting element. The other terminal of the series circuit is connected to a power supply 15 or a substrate potential. The transistors 1 and 2 are driven by complementary signals 114 and 115, respectively. Assuming an approximate circuit, such as shown in Fig. 7, of this circuit configuration in Fig. 13, if synthetic inductance L 1 + L 2 + L 3 of the parasitic inductances 106, 107, and 108 is again replaced with L 3 , this L 3 corresponds to L 3 indicated by the arrow in Fig. 7. In addition, the parasitic inductance 112, i.e., L 4 in Fig. 5 illustrating the prior art, can be set to nearly 0, since the CR series circuit is formed in the integrated circuit. That is, the parasitic inductance produced in the prior art between the collectors of the bipolar transistors for driving the semiconductor light emitting element and the CR series circuit can be decreased to 0. In other words, in the case of this prior art L 1 = L 2 = 0 is accomplished. In the previously mentioned case, Equation 14 yields Equation 15 below because L 4 = 0: 1/√(L 3 C) = R/2L 3 For this reason, by properly choosing the resistance, R, of the resistor 109 and the capacitance, C, of the capacitor 110 in association with the parasitic resistance L 3 , the current flowing through the semiconductor light emitting element when complementary pulse signals are applied to the semiconductor light emitting element driving circuit can be set in a critical state which is between the state in which the current varies in an oscillating manner and the state in which the current varies in an unoscillating manner. Second Embodiment In this embodiment, the constant current circuit denoted by reference numeral 30 in Fig. 13 is constituted by MOS transistors such as shown in Fig. 12. In the semiconductor light emitting element driving circuit in each of the first and second embodiments as described above, a snubber circuit, which consists of a resistor 109, and a capacitor 110 in series, is provided in the same integrated circuit in which the differentially connected bipolar transistors 1 and 2 are formed, and is directly connected to the collectors of the bipolar transistors in the integrated circuit. Therefore, the parasitic inductance that is produced in conventional circuits, which use an external snubber circuit, is not generated between the collectors of the bipolar circuits for driving the semiconductor light emitting element and the series circuit. Consequently, by properly choosing the resistance of the resistor 109 and the capacitance of the capacitor 110 in association with the parasitic resistance occurring outside the circuit, the current flowing through the semiconductor light emitting element when complementary pulse signals are applied to the semiconductor light emitting element driving circuit can be set in a critical state which is between the state in which the current varies in an oscillating manner and the state in which the current varies in an unoscillating manner. With this setting, it is possible to prevent overshoot and perform high-speed switching in the semiconductor light emitting element driving circuit. Fig. 14 is a sectional view showing a portion of an IC chip in which a driving circuit, embodying the present invention, is formed. In Fig. 14, only one bipolar transistor BPT and two MOS transistors MOS are illustrated, and other components such as a protective layer are omitted. The parts shown in Fig. 14 are a p-type silicon substrate 201, an n + -type collector buried layer 202, a p-type well 203, an n - -type epitaxial layer 204, a p-type base 205, and an n + -type emitter 206. The two NMOS transistors MOS have sources/drains 207 and 208 and gates 210. A field insulating film 209 isolates the elements. A collector line 211 of the bipolar transistor is connected to the terminal of a light emitting element. A base line 212 serves as an input terminal, and a line 213 connects a constant current source to the bipolar transistor. A ground line 214 is connected to a low-potential reference voltage source. Fig. 15 is a sectional view showing a portion of an IC chip in which a snubber circuit of a resistor R and a capacitor C, in series, is formed. In this snubber circuit in Fig. 15, n + -type diffusion layers 221 and 222 are formed, and a series connecting line 215 connects the resistor R and the capacitor C. The circuit also includes an insulating interlayer 230. The driving circuit as illustrated in Fig. 14 may be formed into a single chip by a fabrication method called a BiMOS process or a BiCMOS process. If necessary, the series circuit shown in Fig. 15 may be also monolithically integrated with the driving circuit. Figs. 16A and 16B illustrate examples of systems using the circuit of the present invention. Fig. 16A shows a printer, and Fig. 16B shows an optical communication system. The printer shown in Fig. 16A comprises a photosensitive drum DRM, a cleaner CLN, a charger CGR, and an exposure unit EXP which employs the driving circuit of the present invention, a developing unit DVLP, and a recording medium P. The exposure unit EXP uses an LED array or a laser diode as a light emitting element and forms latent images on the photosensitive drum by using light from the light emitting element. In the communication system shown in Fig. 16B, the transmission side includes a laser diode LD as a light emitting element and a transmitter SYS1 with a driving circuit, and the reception side includes a photodiode sensor PHD and a receiver SYS2. An optical fibre OFR also is provided. The driving circuit is adopted in the transmitter SYS1.

Claims: A driving circuit for driving a light emitting element (4), which circuit comprises: a pair of differentially connected bipolar transistors (1,2) integrated in a semiconductor chip (117); a constant current source (30) connected to the emitter of each said bipolar transistor; input means (17,18), connected to the base of each said bipolar transistor for applying complementary control signal pulses (114, 115) thereto; a first resistor (5) connected at one end to the collector of one (2) of said bipolar transistors (1, 2); and a snubber circuit (109, 110) consisting in a second resistor (109) and a capacitor (110) connected in series therewith, connected at one end to the collector of the other one (1) of said bipolar transistors (1, 2), and connected at its other end to the semiconductor chip substrate or to a power line (16) of said semiconductor chip substrate; characterised in that : said snubber circuit (109, 110) also is integrated in the semiconductor chip (117). A driving circuit according to claim 1 wherein said constant current source (30) and said first resistor (5) also are integrated in said semiconductor chip (117). A driving circuit according to either of claims 1 or 2, wherein said constant current source (30) comprises: a current source (51); a voltage line (55) connected to said current source (51); a ground line (16); a MOS transistor (53) connected between said voltage line (55) and said ground line (16), with its gate also connected to said voltage line (55); a current line (57) connected to the emitter of each said bipolar transistor (1, 2); an array of MOS transistors (31, 32, ... 3n) connected in parallel between said current line (57) and said ground line (16); and time constant circuit components, namely a respective resistor (41, 42, ... 4n) connected between said voltage line (55) and the base of each respective array transistor (31, 32, ... 3n), and a respective capacitor (131, 132, ... 13n) connected between the base of each respective array transistor (31, 32, ... 3n) and said ground line (16). A driving circuit, as claimed in any preceding claim, together with the light emitting element (4), which latter is, at one end, connected to said first resistor (5) via a voltage line (15), and, at its other end, connected to the collector of said other one (1) of said bipolar transistors (1,2). A driving circuit together with a light emitting element as claimed in claim 4, wherein said light emitting element (4) also is integrated in said semiconductor chip (117).

Inventors: NAKAMURA HIROYUKI, NAKAMURA, HIROYUKI
Applicants: CANON KK, CANON KABUSHIKI KAISHA
Application year: 1995
Main classification: H03K17
IPC codes: H03K17