Voltage regulator for an output driver with reduced output impedance

Abstract: An output driver circuit for an integrated circuit is disclosed, where the output driver drives an output terminal with a high logic level having a voltage limited from the power supply voltage of the integrated circuit. The limited voltage is provided by applying a limited output high voltage to an output buffer, such that the drive signal applied to the gate of the pull-up transistor in the output driver is limited by the limited output high voltage applied to the output buffer. A voltage reference and regulator circuit for generating the limited output high voltage, or for generating a reference voltage for application to a circuit other than an output buffer and that demands sink current, is also disclosed. The voltage reference and regulator is based on a current mirror, in which the sum of the current in the current mirror is controlled by a bias current source which may be dynamically controlled within the operating cycle or programmed by way of fuses. A sink current path circuit, according to various alternatives, is disclosed as providing an additional sink current path in the event that the limited output high voltage, or reference voltage, exceeds the desired level.

Description: This invention is in the field of integrated circuits, and is more particularly directed to output drive circuitry thereof. In modern digital integrated circuits, particularly those fabricated according to the well-known complementary metal-oxide-semiconductor (CMOS) technology, circuit operation often depends upon the availability of a stable reference voltage. For example, many functional circuits internal to an integrated circuit rely upon current sources that conduct a stable current. Examples of such functional circuits include differential amplifiers, current mirrors, operational amplifiers, level shift circuits, and circuits that themselves generate reference voltages. Since current sources are generally implemented by way of n-channel field effect transistors, the stability of the current source depends upon the stability of the reference voltage applied to the gate of the n-channel field effect transistor. Referring now to Figure 11, a conventional voltage reference generator circuit based on a current mirror is illustrated. The voltage reference circuit of Figure 11 includes p-channel transistors 1, 3 connected in a current mirror fashion, with their sources biased to V cc and their gates connected together at the drain of transistor 1. The drain of transistor 1 is connected to the drain of n-channel transistor 7, which has its gate connected to receive a fraction of V cc determined by voltage divider 5. Transistors 1 and 7 constitute the reference leg of the current mirror. The drain of transistor 3 is connected to the drain and gate of an n-channel transistor 9, at which the reference voltage V ref is produced. Transistors 3 and 9 thus constitute the mirror leg of the current mirror, as the current conducted by transistor 3 "mirrors" that conducted by transistor 1. The sources of transistors 7, 9 are connected together to current source 11, which conducts a current i BIAS that, in this example, is the sum of the currents through the reference and mirror legs. This conventional current mirror-based voltage reference circuit of Figure 11 is beneficial in certain applications. For example, if transistors 3 and 9 are made to be quite large (i.e., channel width to channel length ratio being relatively large), the output impedance of the circuit of Figure 11 will be quite low, allowing the circuit to source and sink relatively large currents without significant modulation in the voltage at line V ref . If such large source and sink currents are to be provided, however, it is preferable that the mirror ratio (i.e., ratio of the size of transistor 3 to that of transistor 1) be large, to reduce the DC current drawn by the circuit and to provide current amplification. However, it has been observed that the circuit of Figure 11 is subject to development of offset voltages therein. For example, if significant source current is required to be produced on line V ref , it has been found that the voltage at the drains of transistors 1 and 7 will drop since the current through transistor 9 is reduced to near zero (all of the current in the mirror leg being sourced to line V ref ). Because of the small size of transistor 1 required for the mirror ratio to be large enough to provide the desired source current, and due to its diode configuration, transistor 1 is not able to rapidly pull up the node at its drain, which will in turn allow the voltage on line V ref to overshoot its desired voltage (transistor 3 being turned on relatively strongly). This overshoot on line V ref is undesirable when applied to circuits that rely upon a stable voltage on line V ref . Other circuits that may receive the voltage on line V ref may place significant sourcing or sinking current demands on line V ref , which will also undesirably affect the stable voltage. In addition, because the current per unit channel width is different in the reference leg relative to the mirror leg, particularly as conducted by transistors 7 and 9, the voltage V ref can settle to an incorrect steady-state value. Still further, it is desirable to provide as much current drive to the output of the voltage reference circuit as possible. However, while the source current provided by transistor 3 may be quite large (limited, of course, by the current mirror ratio), the sink current that may be conducted by transistor 9 is limited to the value i BIAS conducted by current source 11. As such, the output impedance for the sink situation will generally be limited by the value of current i BIAS that can be tolerated from a power dissipation standpoint. EP-A-0427557 discloses an amplifier circuit in which a current mirror circuit is used in a three-stage amplifier circuit which is designed to protect the output current of the amplifier exceed an excessive value by dropping its output voltage to zero following an excessively high input voltage. To do this a differential amplifier based upon a current mirror is used, and is followed by a current limiting stage which controls the current drawn by an output stage transistor supplying current to an output terminal. IBM Technical Disclosure Bulletin, vol. 32, no. 10a, April 1990, pages 26-28, entitled "On-chip voltage regulators with improved ripple rejection", discloses a voltage regulator device in which an additional reference level generator is used to power a differential amplifier which controls the series regulating element operating in the source-follower mode and thereby reduces the ripple noise coupling from the external supply to the regulated supply. In accordance with a first aspect of the invention, there is provided a voltage reference circuit, comprising: a first reference transistor, having a source/drain path connected on a first end to a power supply voltage, and having a gate connected to a drain end of its source/drain path; a second reference transistor having a source/drain path connected between a second end of the source/drain path of the first reference transistor and a common node, and having a gate for receiving a reference voltage; a first mirror transistor, having a source/drain path connected between the power supply voltage and an output node, and having a gate connected to the gate of the first reference transistor; a second mirror transistor, having a source/drain path connected between the output node and the common node, and having a gate; a bias current source, coupled to the common node, for conducting a current including the sum of the currents in the second reference transistor and the second mirror transistor; a sink current path circuit, connected to the output node, for conducting current responsive to the voltage at the output node exceeding a desired level, wherein the sink current path circuit comprises: a sink transistor having a conduction path connected between the output node and ground, and having a control electrode; a sink bias circuit coupled to the output node and to the control electrode of the sink transistor, for turning on the sink transistor responsive to a rise in the voltage at the output node. The circuit may comprisea a voltage divider, for generating the reference voltage applied to the gate of the second reference transistor and to the gate of the second sink reference transistor. In accordance with a second aspect of the invention, there is provided an output driver circuit for an integrated circuit, comprising: a pull-up drive transistor, having a conduction path coupled between a power supply voltage and an output terminal, and having a control electrode; circuitry for limiting the drive of the pull-up drive transistor responsive to a bias voltage; and a voltage reference circuit according to the first aspect and including the above-referenced voltage divider, for producing the bias voltage, wherein the reference voltage generated by the voltage divider is based upon the power supply voltage, the voltage reference circuit further comprising a reference leg connected between the first reference transistor, the second reference transistor and the common node, and a mirror leg connected between the first mirror transistor, the second mirror transistor and the common node, said reference leg receiving the divided voltage from the voltage divider and conducting a reference current responsive thereto, said mirror leg conducting a mirrored current responsive to the reference current and deriving, at an output, the bias voltage based upon said mirrored current; wherein the bias current source is coupled to the reference leg and the mirror leg of the current minor, for controlling the reference current and the mirror current conducted thereby; and wherein the sink current path circuit is coupled to the bias voltage output of the current mirror, for conducting sink current responsive to the bias voltage output exceeding a desired level. The invention may be implemented into an integrated circuit as an output driver system in which the voltage applied to the gate of the pull-up device is set at a voltage above the minimum V OH operating window but below the circuit power supply. The system includes circuitry for generating the proper gate voltage to the pull-up device based upon a regulated voltage. The regulated voltage is produced by a combination voltage reference and voltage regulator circuit, where the regulated voltage is shifted by a threshold voltage to allow the output driver to be an n-channel device with its body node back-biased. The regulated voltage serves as the bias voltage in the output buffer, so that the output voltage of the output buffer as applied to the gate of the n-channel pull-up device is limited. According to another embodiment of the invention, circuitry is provided to assist in the sinking of current from the output of the voltage reference and voltage regulator circuit. This circuitry provides such additional sink current capability without requiring the bias current source in the voltage reference and voltage regulator circuit to conduct additional current. As a result, the output impedance of the voltage reference and regulator circuit is reduced on the sink side without increasing the power dissipated by the circuit. The present invention may also be applied to other circuitry, besides output drivers, that require a stable voltage reference. Embodiments of the invention will be described by way of example only, in the following description with reference to the accompanying drawings, in which: Figure 1 is an electrical diagram, in block form, of an integrated memory circuit incorporating output drive circuitry according to the preferred embodiment of the invention. Figure 2 is an electrical diagram, in block form, of the output drive circuitry according to the preferred embodiment of the invention. Figure 3 is an electrical diagram, in schematic form, of a voltage reference and regulator circuit according to the preferred embodiment of the invention. Figure 4 is an electrical diagram, in schematic form, of a bias current source as used in the voltage reference and regulator circuit according to the preferred embodiment of the invention. Figures 5 and 6 are timing plots of the operation of the voltage reference and regulator circuit according to the preferred embodiment of the invention in the absence and presence, respectively, of an offset compensating current. Figure 7 is an electrical diagram, in schematic form, of a dynamic bias control circuit as used in the voltage reference and regulator circuit according to the preferred embodiment of the invention. Figure 8 is a timing diagram illustrating the operation of the circuit of Figure 7 in an integrated circuit memory. Figure 9 is an electrical diagram, in schematic form, of a bias current source according to an alternative embodiment of the invention, including programmable bias current levels. Figure 10 is an electrical diagram, in schematic form, of a voltage reference and regulator circuit according to an alternative embodiment of the invention. Figure 11 is an electrical diagram, in schematic form, of a voltage reference circuit according to the prior art. Figure 12 is an electrical diagram, in schematic form, of a voltage reference and regulator circuit according to an alternative embodiment of the invention, including circuitry for providing an additional sink current path. Figure 13 is an electrical diagram, in schematic form, of a voltage reference and regulator circuit including circuitry for providing an additional sink current path, according to another alternative embodiment of the invention. As will become apparent from the following description, it is contemplated that the present invention may be implemented into many types of integrated circuits that generate digital output signals. Examples of such integrated circuits include memory circuits of the read-only, programmable read-only, random access (either static or dynamic), and FIFO types, timer circuits, microprocessors, microcomputers, microcontrollers, and other logic circuits of the general or programmable type. For purposes of description, the preferred embodiment of the invention will be described for the example of a memory integrated circuit, as memory circuits are contemplated to be often used to provide output data to an integrated circuit (such as a microprocessor) having a lower power supply voltage. Figure 1 illustrates a block diagram of read/write memory 10 in which the preferred embodiment of the present invention is implemented. Memory 10 includes a plurality of memory cells arranged in memory array 16. In general, memory 10 operates to receive an M bit address and, synchronous to a system clock (denoted "CLK"), to output an N bit data quantity. Integers M and N are selected by the designer according to the desired memory density and data path size. Selected memory cells in memory array 16 are accessed by operation of address register 12, timing and control circuit 14, and address decoder 17, in the conventional manner and as will be described hereinbelow. Data terminals 28 allow for communication of data to and from read/write memory 10; while data terminals 28 in this example are common input/output terminals, it will of course be understood that separate dedicated input terminals and output terminals may alternatively be implemented in memory 10. Data is read from the selected memory cells in memory array 16 via read circuitry 19 (which may include sense amplifiers, buffer circuitry, and the like, as conventional in the art), output buffers 21, and output drivers 20; conversely, data is written to the selected memory cells in memory array 16 via input drivers 18 and write circuitry 17. Address register 12 includes an integer M number of address inputs labeled A 1 through A M . As known in the memory art, the address inputs allow an M bit address to be applied to memory 10 and stored in address register 12. In this example, memory 10 is of the synchronous type, and as such the address value at address inputs A is clocked into address register 12 via CLK, where CLK is passed to address register 12 from timing and control circuit 14. Once the address is stored, address register 12 applies the address to memory array 16 via address decoder 17, in the usual manner. Timing and control circuit 14 is also illustrated as having a generalized set of control inputs (denoted "CTRL") which is intended to represent various control and/or timing signals known in the art, such as read/write enable, output enable, burst mode enable, chip enable, and the like. In this example, memory 10 receives electrical power from power supply terminal V cc , and also has a reference voltage terminal GND. According to the preferred embodiment of the invention, memory 10 will be presenting output data at data terminals 28 for receipt by another integrated circuit that is powered by a power supply voltage lower than that applied to terminal V cc of memory 10. For example, the power supply voltage applied to terminal V cc of memory 10 may nominally be 5 volts (relative to the voltage at terminal GND) while an integrated circuit receiving data presented by memory 10 at terminals 28 may have a power supply voltage of nominally 3.3 volts. In order to allow this condition, the maximum voltage driven by output drivers 20 of memory 10 at data terminals 28 must be at or near this lower power supply voltage (i.e., at or near 3.3 volts), to avoid damage to the downstream integrated circuit. As will be described in detail hereinbelow, the preferred embodiment of the present invention is intended to provide such limitation on the maximum output high level voltage driven by output drivers 20 of memory 10. Memory array 16 is a standard memory storage array sized and constructed according to the desired density and architecture. In general, array 16 receives decoded address signals from address decoder 17, responsive to which the desired one or more memory cells are accessed. One of the control signals, as noted above, selects whether a read or write operation is to be performed. In a write operation, input data presented to data terminals 28, and communicated via input buffers 18, are presented to the selected memory cells by write circuitry 21. Conversely, in a read operation, data stored in the selected memory cells are presented by read circuitry 19 to output buffers 21. Output buffers 21 then produce control signals to output drivers 20, to present digital output data signals at data terminals 28. In either case, internal operation of memory 10 is controlled by timing and control circuitry 14, in the conventional manner. According to the preferred embodiment of the invention, memory 10 further includes output buffer bias circuit 22. Output buffer bias circuit 22 generates a bias voltage on line VOHREF that is presented to output buffers 21 so that the control signals presented by output buffers 21 in turn limit the maximum output voltage driven by output drivers 20 on data terminals 28. As indicated in Figure 1, and as will be described in further detail hereinbelow, output buffer bias circuit 22 according to the preferred embodiment of the invention is controlled by timing and control circuitry 14 according to the timing of the memory access cycle. Referring now to Figure 2, the construction of output buffer bias circuit 22 and its cooperation with output buffers 21 and output drivers 20 according to the preferred embodiment of the present invention will be described in further detail. As shown in Figure 2, output buffer bias circuit 22 includes voltage reference and regulator 24, which produces a regulated voltage VOHREF at its output. Output buffer bias circuit 22 further includes bias current source 26 which, as will be described in further detail hereinbelow, is controlled by a clock signal generated on line C50 by timing and control circuitry 14; bias current source 26 produces a bias current i BIAS used by voltage reference and regulator 24 in generating the voltage on line VOHREF. Also according to this embodiment of the invention, voltage reference and regulator 24 receives an offset compensating current i NULL from offset compensating current source 28. Output buffer bias circuit 22 further includes V t shift circuit 30, which serves to set the voltage VOHREF. The detailed construction and operation of output buffer bias circuit 22 and its respective constituent blocks will be described in further detail hereinbelow. As shown in Figure 2, output buffer bias circuit 22 also includes sink current path circuitry 25, according to certain alternative embodiments of the invention to be described in further detail hereinbelow. Sink current path circuitry 25, in providing an additional current path between ground and line VOHREF at the output of output buffer bias circuit 22, provides a reduction in the output impedance or output buffer bias circuit 22 in the condition when significant sink current is required to be conducted thereby. As such, overshoot of the voltage on line VOHREF as a result of sink current requirements, for example output driver switching, will also be minimized. This output impedance reduction is obtained without increasing the value of current i BIAS required to be conducted by bias current source 26, and thus without increasing the DC power dissipation of output buffer bias circuit 22. Voltage VOHREF is presented to each of the output buffers 21. As such, output buffer bias circuit 22 serves multiple ones of output buffers 21; in many cases, depending upon the number of output buffers 21, a single output buffer bias circuit 22 may suffice to control all of the output buffers 21. Each output buffer 21 receives complementary data inputs DATA, DATA*, which are generated by read circuitry 19 (see Figure 1). For example, output buffer 21 j receives complementary data inputs DATA j , DATA j * (the * indicating logical complement). Each output buffer 21 presents control signals (shown as PU and PD for output buffer 21 j ) to a corresponding output driver 20. Each output driver 20 drives a corresponding data terminal 28. While, as shown in Figure 1, data terminals are common input/output terminals, the input side (i.e., data input buffers, etc.) are not shown in Figure 2 for the sake of clarity. Each output buffer 21 in this embodiment of the invention is implemented as an n-channel push-pull driver. Referring specifically to output driver 20 j , which is shown in detail in Figure 2 (it being understood that the other output drivers 20 are similarly constructed), n-channel pull-up transistor 32 has its drain biased to V cc and its source connected to data terminal 28 j , and n-channel pull-down transistor 34 has its drain connected to data terminal 28 j and its source biased to ground Output drivers 20 also preferably include electrostatic discharge protection devices (not shown), as is conventional in the art. The gates of transistors 32, 34 receive control signals PU, PD, respectively, from output buffer 21. As will be appreciated by those of ordinary skill in the art, since V cc (nominally 5 volts, for example) biases the drain of pull-up transistor 32, the voltage of line PU applied to the gate of transistor 32 must be properly controlled to ensure that the maximum voltage to which transistor 32 drives data terminal 28 j in presenting a logical one (referred to as V OH maximum) does not exceed the limit (e.g., 3.3 volts). The way in which this limitation is accomplished according to the preferred embodiment of the invention will be described hereinbelow. As is shown in Figure 2, the body node of n-channel pull-up cransistor 32 is preferably biased to ground, rather than to its source at data terminal 28 j . It will be appreciated by those of ordinary skill in the art that this body node bias for n-channel pull-up transistor 32 is preferred to avoid vulnerability to latchup. However, as will also be appreciated, this bias condition for transistor 32 will effectively increase its threshold voltage, making it more difficult to limit V OH maximum driven by output driver 20. This difficulty is due to the higher voltage to which line PU must be driven in order tc turn on transistor 32. The preferred embodiment of the present invention, as will be described hereinbelow, addresses this difficulty in such a way as to allow the body node of transistor 32 to be back biased (i.e., to a voltage other than that of its source). Output buffer The construction of output buffer 21 j as shown in Figure 2 will now be described in detail, it being understood that the other output buffers 21 are similarly constructed. Output buffer 21 j receives the data input lines DATA j , DATA j * at an input of respective NAND functions 40, 42. Output enable line OUTEN is also received at an input of each of NAND functions 40, 42 to perform an output enable function as will be described hereinbelow. The output of NAND function is applied to the gates of p-channel transistor 36 and n-channel transistor 38. P-channel transistor 36 has its source biased to the voltage VOHREF generated by output buffer bias circuit 22, and has its drain connected to line PU. N-channel transistor 38 has its drain connected to line PU and its source biased to ground. As such, transistors 36, 38 form a conventional CMOS inverter for driving line PU with the logical complement of the logic signal presented by NAND function 40. However, the high voltage to which line PU is driven by transistor 36 is limited to the voltage VOHREF generated by output buffer bias circuit 22. Since line PU is presented to the gate of n-channel pull-up transistor 32 in output driver 20 j , the voltage VOHREF thus will control the maximum drive of pull-up transistor 32, and thus the voltage to which data terminal 28 j is driven. On the low side, the output of NAND function 42 is applied to the input of inverter 43 (which, in this case, is biased by V cc ). The output of inverter 43 drives line PD, which is applied to the gate of n-channel pull-down transistor 34. In operation, with output enable line OUTEN at a high logic level, the state of NAND functions 40, 42 are controlled by the state of data input lines DATA j , DATA j *, and will be the logical complement of one another (since data input lines DATA j , DATA j * are the logical complement of one another). A high logic level on line DATA j will thus result in a low logic level at the output of NAND function 40, turning on transistor 36 so that the voltage VOHREF is applied to the gate of transistor 32 via line PU, driving data terminal 28 j to a high logic level (limited by the voltage of VOHREF as noted above); the output of NAND function 42 in this condition is high (data line DATA j * being low) which, after inversion by inverter 43, turns off transistor 34 in output driver 20 j . In the other data state, the output of NAND function 40 will be high (data line DATA j being low), turning on transistor 38 to pull line PU low to turn off transistor 32; the output of NAND function 42 will be low, causing inverter 43 to drive line PD high and turn on transistor 34, pulling data terminal 28 j low. With output enable line OUTEN at a low logic level, the outputs of NAND functions 40, 42 are forced high regardless of the data state applied by data input lines DATA j , DATA j *; as a result, transistors 32, 34 are both turned off, maintaining data terminal 28 j in a high impedance state. As noted above, the voltage on line VOHREF in this embodiment of the invention determines the drive applied to n-channel pull-up transistors 32 in output drivers 20. According to this embodiment of the invention, therefore, the construction of output buffer 21 in providing the voltage VOHREF to the gate of pull-up transistor 32 is particularly beneficial, as it is implemented with a minimum of transistors, and can rapidly switch to effect fast transitions at data terminals 28. In addition, no series devices are required in output drivers 20 to limit V OH maximum according to this embodiment of the invention such series devices necessarily reducing the switching speed of output drivers 20 and also introducing vulnerability to electrostatic discharge and latchup. Furthermore, no boocstrapping of the gate drive to n-channel transistor 32 is required according to this embodiment of the invention, thus avoiding voltage slew and bump sensitivity. The construction of output buffer bias circuit 22 in presenting the proper voltage VOHREF, so that memory 10 in this embodiment of the invention may drive a logic high level to a safe maximum level for receipt by integrated circuits having lower power supply voltages will now be described in detail, with respect to each of the circuit functions of output buffer bias circuit 22 shown in Figure 2. Voltage reference and regulator with V t shift Referring now to Figure 3, the construction and operation of voltage reference and regulator 24 will now be described in detail, in cooperation with the other elements of output buffer bias circuit 22. As shown in Figure 3, voltage reference and regulator 24 is constructed in current mirror fashion. P-channel transistors 44 and 46 each have their sources biased to V cc , and have their gates connected together. In the reference leg of this current mirror, the drain of transistor 44 is connected to its gate, and to the drain of n-channel transistor 48. The gate of n-channel transistor 48 is connected to a voltage divider constructed of resistors 47, 49 connected in series between V cc and ground, where the gate of transistor 48 is connected at the point between resistors 47 and 49 to receive the desired fraction (e.g., 60%) of the V cc power supply voltage. Alternatively, each leg of the resistor divider may be constructed of a series of resistors that are initially shorted out by fuses; opening of selected fuses can thus allow programmability of the voltage applied to the gate of transistor 48. The source of transistor 48 is connected to bias current source 26. In the mirror leg of this current mirror, the drain of transistor 46 is connected, at output node VOHREF, to the drain of n-channel transistor 50. The gate of transistor 50 is coupled to node VOHREF via V t shift circuit 30, in a manner that will be described in further detail hereinbelow. The source of n-channel transistor 50 is connected to the source of transistor 48 in the reference leg and thus to bias current source 26. As noted above, bias current source 26 conducts a current i BIAS , which will be the sum of the currents in the reference and mirror legs in the current mirror of voltage reference and regulator 24 (i.e., the sum of the currents through transistors 48 and 50). The current i BIAS is primarily produced by n-channel transistor 52 which has its drain connected to the sources of transistors 48 and 50, its source biased to ground, and its gate controlled by bias reference circuit 54. As will be further described in detail below, according to the preferred embodiment of the invention, dynamic bias circuit 60 is also provided for controlling the current i BIAS to be decreased at certain times in the memory access cycle (under the control of clock signal C50), to optimize the output impedance of voltage reference and regulator 24 for different portions of the memory access cycle. V t shift circuit 30 provides the bias of the gate of n-channel transistor 50 in the mirror leg of voltage reference and regulator 24 in this preferred embodiment of the invention, to ensure that voltage VOHREF is shifted upward by an n-channel threshold voltage, considering that voltage VOHREF will be applied (via output buffers 21) to the gate of n-channel pull-up transistors 32 in output drivers 21. The way in which this shift is effected will be described hereinbelow with the operation of voltage reference and regulator 24. The operation of voltage reference and regulator 24 will now be described in detail, at a point in the memory cycle during which output data is to be presented at data terminals 28. Bias reference circuit 54 presents a bias voltage to the gate of n-channel transistor 52 to set the value of i BIAS conducted through the current mirror; dynamic bias circuit 60 is effectively off at this time. The divided voltage generated by resistors 47, 49, which is presented as a reference voltage to the gate of n-channel transistor 48, determines the extent to which transistor 48 is conductive, and thus determines the bias condition at the drain of p-channel transistor 44. The current conducted by transistor 44 is mirrored by transistor 46 in the mirror leg, and will thus be a multiple of the current conducted by transistor 44 (as will be discussed hereinbelow). The voltage VOHREF at the drains of transistors 46, 50 will be determined by the voltage at the drains of transistors 44, 48, by the relative sizes of the transistors in the circuit, and by the effect of V t shift circuit 30. As is well known in the art of current mirror circuits, the gate voltage of transistor 50 will tend to match that at the gate of transistor 48, due to the feedback of the voltage at line VOHREF to the gate of transistor 50, considering the differential amplifier effect of voltage reference and regulator 24. V t shift circuit 30, however, includes transistor 56, connected in diode fashion with its gate connected to its drain at VOHREF, and with its source connected to the gate of transistor 50, so that a threshold voltage drop is present between line VOHREF and the gate of transistor 50. Transistor 56 is constructed similarly as -one of n-channel pull-up transistors 32 in output drivers 20, particularly in having the same or similar gate length and in having the same body node bias (e.g., to ground). N-channel transistor 58 has its drain connected to the source of transistor 56, and has its gate controlled by bias reference circuit 54, to ensure proper current conduction through transistor 56 so that an accurate threshold voltage drop is present across transistor 56. As a result of V t shift circuit 30, the voltage at line VOHREF will be boosted from the reference voltage at the gate of transistor 48 by a threshold voltage value that closely matches the threshold voltage of the n-channel pull-up transistor 32 of output drivers 20. This additional threshold voltage shift is necessary considering that the voltage VOHREF will be applied to the gate of an n-channel pull-up transistor 32 in output drivers 20, thus ensuring adequate high level drive. The V t shift is effected by circuit 30 in a way that does not increase the output impedance of voltage reference and regulator 24, particularly in the impedance to sink current through transistor 50 in the event of fluctuations of voltage VOHREF caused by switching output buffers 21. The implementation of circuit 30 also introduces minimum offset voltage into voltage reference and voltage regulator 24, and requires only two additional transistors 56, 58 without adding an entire stage. It is of course contemplated that the voltage generated on line VOHREF by voltage reference and regulator 24 may be applied to control the logic level high drive of output driver 20 in alternative ways to that described hereinabove relative to the preferred approach of controlling the source voltage of pull-up transistors 36 in output buffers 21. For example, the voltage generated on line VOHREF may be directly applied to the gate of a transistor in series with the pull-up transistor in output driver 20 or, in another example, the voltage generated on line VOHREF may be applied to the gate of a transistor in series with the pull-up transistor in output buffer 21; in each of these alternative cases, the reference voltage on line VOHREF limits the drive applied to the output terminal. In such alternatives, however, one of ordinary skill in the art will recognize that the absolute level of the reference voltage on line VOHREF may have to be shifted from that utilized in the foregoing description. Offset compensating current source It is desirable for voltage reference and regulator 24 to have extremely low output impedance, so that substantial current may be sourced to or sinked from line VOHREF without significant modulation of the voltage on line VOHREF. As noted above, since the voltage on line VOHREF controls the maximum output high level voltage V OH maximum so as not to damage an integrated circuit receiving the output logic signals at data terminals 28 while still providing the maximum output drive, it is important that the voltage on line VOHREF remain steady near the regulated level. In voltage reference and regulator 24, therefore, it is desirable that the drive capabilities, and thus the transistor sizes (i.e., ratio of channel width to channel length, or W/L) of transistors 46 and 50 be quite large. This large size for transistors 46, 50 will allow voltage reference and regulator 24 to rapidly source current (from V cc through transistor 46 to line VOHREF) or sink current (from line VOHREF through transistors 50, 52 to ground). For example, the W/L of transistor 46 may be on the order of 1200, the W/L of transistor 50 may be on the order of 600, and the W/L of transistor 48, in this example, may be on the order of 300. In addition, it is desirable that the W/L of transistor 46 be larger than that of transistor 44, so that a sizable mirror ratio may be obtained, thus increasing the source current available on line VOHREF; further, it is desirable that the W/L of transistor 48 be significantly larger than that of transistor 44, for high gain. In the above example, the W/L of transistor 44 may be on the order of 60, in which case the mirror ratio of voltage reference and regulator 24 would be on the order of 20. The maximum source current i source will be determined as follows: i source max = i BIAS ( W L ) 46 ( W L ) 44 In the above example, the maximum source current i source max will be on the order of 20 times i BIAS The maximum sink current of voltage reference and regulator 24 will be equal to i BIAS , which is controlled by bias current source 26. In this embodiment of the invention, it will of course be appreciated that the source current will be the more critical parameter for this embodiment of the invention, as it controls the turn-on of pull-up transistors 32 in output drivers 21. However, since the currents through the reference and mirror legs of voltage reference and regulator 24 are not equal to one another, an offset voltage can develop between the nodes at the drains of transistors 44, 48, on one hand, and the drains of transistors 46, 50, on the other hand. This offset voltage can be on the order of 300 to 400 mV, and will increase with increasing i BIAS . Furthermore, since the W/L of transistor 48 is substantially larger than that of transistor 44 and due to the diode configuration of transistors 44 (gate tied to drain), transistor 44 is unable to rapidly pull the voltage at the drain of transistor 48 (and the gates of transistors 44, 46) high when necessary. For example, when multiple ones of output drivers 21 simultaneous switch on their respective pull-up transistors 32, substantial source current from voltage reference and regulator 24 is required to maintain the voltage on line VOHREF at the proper level. This source current tends to initially pull down the voltage on line VOHREF, which in turn will pull down the voltage at the drains of transistors 44, 48 in the reference leg of voltage reference and regulator 24, since transistor 48 will be required to temporarily supply most of the current i BULK required by current source 26 because virtually all of the current conducted by transistor 46 is directed to line VOHREF. However, because of its relatively small size (for high mirror ratio), transistor 44 is unable to rapidly pull up the voltage at its drain by itself; if this voltage remains low, once the transient demand for source current is over, the voltage VOHREF will overshoot its steady state voltage, because transistors 44 and 46 will be turned on strongly by the low voltage at their gates. As discussed above, overshoot of the voltage VOHREF can damage downstream integrated circuits that have lower power supply voltages. According to the preferred embodiment of the invention, therefore, offset compensating current source 29 is provided, to source current i NULL into voltage reference and regulator 24 at the drains of transistors 44, 48. The size of bias current source transistor 52 must therefore be adequate to conduct the additional current i NULL that will be provided into the reference leg of voltage reference and regulator 24 beyond the current mirror; of course, an additional transistor may be provided in parallel with transistor 52 to conduct this additional current. The current i NULL is intended to equate the current per unit channel width conducted by transistor 48 with the current per unit channel width conducted by transistor 50, so that no offset voltage results, as well as easing the load of transistor 48 on transistor 44, and allowing the voltage at the drains of transistors 44 and 48, and thus at the gates of transistors 44, 46, to be rapidly pulled high when necessary. Overshoot of the voltage on line VOHREF is thus prevented. Referring now to Figure 4, the construction of offset compensating current source 28 will be described in detail. In this particular embodiment of the invention, offset compensating current source 28 is controlled by bias reference circuit 54 in bias current source 26 to minimize the number of transistors required for implementation; of course, offset compensating current source may have its own bias reference network, if desired. Bias reference circuit 54 is implemented by way of p-channel transistor 62 having its source biased to V cc and its gate biased by a reference voltage PVBIAS which may be generated by a conventional voltage reference circuit and used elsewhere in memory 10, or which is preferably generated by a compensating bias voltage reference circuit as described in US-A-5 568 084 entitled "Circuit for Providing a Compensated Bias Voltage", assigned to SGS-Thomson Microelectronics, Inc., and incorporated herein by this reference. N-channel transistor 54 is connected in diode fashion, with its gate and drain connected co the drain of transistor 64. The sizes of transistors 62 and 64 are selected to ensure that p-channel transistor 62 remains in saturation for the specified voltage PVBIAS. For example, for a voltage PVBIAS of approximately 2 volts, transistors 62 and 64 with W/L ratios of approximately 15 will maintain transistor 52 in saturation where V cc is nominally 5 volts. The common node at the drains of transistors 62, 64 presents a reference voltage ISVR that is applied to the gate of transistor 52 in bias current source 26, and to offset compensating current source 28. Because of the large currents conducted in voltage reference and regulator 24, as well as the large variations in process parameters and power supply voltages expected over temperature, it is desirable that the operation of bias reference circuit 54 be as stable as possible. The construction of bias reference circuit 54 shown in Figure 4 provides such stability. In the above example, simulation results indicate that the ratio of maximum to minimum current conducted by transistor 52 in bias current source 26, using bias reference circuit 54 to set the gate voltage at node ISVR, over variations in temperature, process parameters, and power supply voltage, is approximately 1.17. Offset compensating current source 28 according to this embodiment of the invention is implemented by a current mirror circuit, in which the reference leg includes p-channel transistor 66 and n-channel transistor 68. The sources of transistors 66, 68 are biased to V cc and ground, respectively, and their drains are connected together. The gate of n-channel transistor 68 receives the reference voltage at node ISVR from bias reference circuit 54, and the gate of p-channel transistor 66 is connected to the common drain node of transistors 66, 68, and to the gate of p-channel transistor 69 in the mirror leg, in typical current mirror fashion. Transistor 69 has its source biased to V cc , such that its drain current provides the current i NULL . The relative sizes of transistors 66, 69 will, of course, determine the mirror ratio, and thus the current i NULL ; a mirror ratio of on the order of 5 will be typical, to produce a current i NULL of on the order of 2.5 mA. As noted above, enough current capability must be provided for transistor 52 to conduct this additional current i NULL ; preferably, an n-channel transistor is provided in parallel with transistor 52, with its gate controlled by line ISVR, and having a size matching that of the mirror circuit of transistors 66, 68, 69, to conduct the additional i NULL current in a matched fashion. Referring now to Figures 5 and 6, the effect of offset compensating current source 28 on the operation of voltage reference and regulator 24 will now be described, based on simulations. Figure 5 illustrates the operation of voltage reference and regulator 24, in the case where the current i NULL is zero, in other words, as if offset compensating current source 28 were not present. Figure 5 illustrates the voltage VOHREF at the output of voltage reference and regulator 24, the voltage V 44 at the common drain node of transistors 44, 48, and the output voltage DQ on one of data terminals 28. Time t 0 indicates the steady-state condition of these voltages, in the case where all data terminals 28 are driving a low output voltage. In the steady-state, for example, the voltage VOHREF is preferably at 3.3 volts (the lower power supply voltage of an integrated circuit receiving the output data from memory 10) plus an n-channel threshold voltage (considering that pull-up transistor 32 in output driver 20 is an n-channel device). At time t 1 , data terminals 28 begin switching to a new data state; in this example, the worst case condition is that where all (e.g., eighteen) data terminals 28 are to switch from a low logic level to a high logic level. As shown in Figure 5, once this switching begins as indicated by the voltage DQ begins rising, the voltages VOHREF and V 44 dip, due to the significant source current required by output buffers 21 on line VOHREF which pulls its voltage down. The voltage V 44 also drops at this time, since the current through transistor 50 is reduced to near zero (all of the current in the mirror leg being required by output buffers 21), forcing transistor 48 to conduct virtually all of the current i BIAS . This additional conduction by transistor 48 in turn drops the voltage at node V 44 . Time t 2 indicates the end of the output transient, such that the source current demand begins decreasing, allowing the voltage on line VOHREF to rise by operation of voltage reference and regulator 24. However, as noted above, because of the small size and diode configuration of transistor 44 required for the mirror ratio to be large enough to provide the source current required by output buffers 21, the voltage at node V 44 remains low for a significant time, and does not begin to rise (slowly) until time t 3 . So long as the voltage at node V 44 remains below its steady-state value, which maintains transistors 44 and 46 turned on strongly, the voltage at line VOHREF is allowed to rise, and indeed rises past its steady-state value by a significant margin (V os ). This rise in VOHREF past its desired value may then be reflected via output buffers 21 and output drivers 20 onto data terminals 28, indeed to the extent as to cause damage to a lower power supply integrated circuit connected to data terminals 28. Referring now to Figure 6, the operation of voltage reference and regulator 24 for the example where the current i NULL is 2.5 mA is illustrated, based on simulation of the same conditions as that shown in Figure 5, and having the same time scale as Figure 5. As before, the switching occurring at time t, causes the voltages VOHREF and V 44 to drop. However, the additional current i NULL applied to the common drain node of transistors 44, 46 assists in the charging of this node, and as a result the time t 3 at which voltage V 44 begins to rise occurs much sooner after the initial switching time t 1 . Since the voltage V 44 begins to rise so quickly in this case, the voltage VOHREF is not allowed to overshoot its steady-state value by nearly as much, nor for nearly as long a time, as in the case of Figure 5 with i NULL = 0. Damage to low power supply integrated circuits connected to data terminals 28 is thus avoided. Dynamic Control of Bias Current As is evident from the foregoing description, it is desirable that the output impedance of voltage reference and regulator 24 be as low as possible during such times as output buffers 21 and output drivers 20 will be switching the state of data terminals 28. This low output impedance allows for significant source and sink current to be provided by voltage reference and regulator 24, without significant modulation in the voltage VOHREF. However, such low output impedance requires that the DC current through voltage reference and regulator 24 to be significant, thus causing significant steady-state power dissipation and the corresponding increase in temperature, decrease in reliability, and load on system power supplies, all of which are undesirable. Referring now to Figure 7, the construction and operation of dynamic bias circuit 60, in controlling the bias current i BIAS within a memory access cycle, will now be described in detail. Dynamic bias circuit 60 is provided as an optional function in voltage reference and regulator 24, for purposes of reducing the steady-state current drawn thereby. As shown in Figure 7, dynamic bias circuit 60 receives clock signal C50, and applies it to the gate of n-channel transistor 72 via inverter 71. Transistor 72 has its drain connected to node ISVR at the output of bias reference circuit 54 and at the gate of current source transistor 52. The source of transistor 72 is connected to the drain of n-channel transistor 74, which has its gate connected to node ISVR and it source biased to ground. In operation, so long as the clock signal C50 remains high, transistor 72 will be off and dynamic bias circuit 60 will not affect the gate bias of transistor 52 nor the value of the current i BIAS conducted thereby. With clock signal C50 low, however, transistor 72 will be turned on and the voltage at the gate of transistor 52 will be reduced due to transistors 72, 74 pulling node ISVR toward ground and reducing the current conducted thereby. The extent to which the gate bias of transistor 52 is reduced by dynamic bias 60 is determined by the size of transistor 74 relative to the size of transistor 64 in bias reference circuit 54 and relative to the size of transistor 52, as will be apparent to those of ordinary skill in the art. This sizing can be readily determined, considering that the gate-to-source voltage of transistor 74 will be the same as that of transistor 64 in bias reference circuit 54. The drain-to-source voltage of transistor 74 will be less than that of transistor 64, however, by the amount of the drain-to-source voltage of transistor 72 when turned on, which will typically be quite small, for example on the order of 100 mv. With both of transistors 64, 74 in saturation, their drain currents will not be significantly affected by their drain-to-source voltages, and as such transistors 64, 74 may be considered to be in parallel with one another when transistor 72 is turned on. Since the current in transistor 52 mirrors that of transistor 64 (in parallel with transistor 74, when transistor 72 is on), clock signal C50 controls the current i BIAS , which effectively changes the current mirror ration of transistor 64 to transistor 52. For example, in the case where the current i BIAS is to be reduced to 50% of its full value except during output switching, the channel width and channel length of transistors 64 and 74 will be the same, if the channel width and channel length of transistors 64 and 52 are the same, as in this example. With transistor 72 turned off, the current i BIAS will equal the current i 64 through transistor 64 in bias reference circuit 54. With transistor 72 turned on (clock signal C50 low), as noted above, transistors 64, and 74 are effectively in parallel with each other and, in this example, have a channel width chat is effectively twice that of transistor 52. The current mirror ratio is therefore one-half, since: W 52 W 64 + W 74 = 1 2 where W 52 , W 64 , W 74 are the channel widths of transistor 52, 64, 74 (channel lengths assumed to be equal). The sum W 64 + W 74 is the effective channel width of transistors 64 and 74 in parallel with one another. Accordingly, the current i BIAS is reduced by one-half during such time as clock signal C50 is low. Referring now to Figure 8, the operation of dynamic bias circuit 60 and its affect on the bias current i BIAS within a memory access cycle will now be described. Time t 0 illustrates the condition of memory 10 at the end of a previous cycle, in the steady state. Data terminals DQ are presenting the output data value DATA 0 from the prior cycle. Clock C50 is low at this time, since output switching is not occurring. Accordingly, the current i BIAS is at one-half of its maximum value, since transistor 72 (Figure 7) is turned on by inverter 71, placing transistor 74 in parallel with transistor 64 of bias reference circuit 54, and thus reducing the mirror ratio of transistor 52. This reduces the current i BIAS drawn by voltage reference and regulator 24 during times in the memory access cycle in which output switching is not expected, and thus during which only the prior data state (i.e., DATA 0 ) is being maintained. The output impedance of voltage reference and regulator 24 may be relatively high during this time, but the voltage on line VOHREF will be maintained at its correct steady-state level. At time t 1 , a new memory access cycle is initiated by input clock CLK going active; alternatively, for example in a fully static memory, clock CLK may correspond to an edge transition detection pulse generated by detection of a transition at address or data input terminals of the memory. Responsive to the leading edge of clock CLK, clock signal C50 is activated after a selected delay corresponding to a time safely short of the minimum expected read access time of the memory. Once clock signal C50 becomes active at time t 2 , transistor 72 is then turned off by operation of inverter 71. Accordingly, the current mirror ratio of transistor 52 is restored to its maximum value (unity, in this example) prior to such time as the output buffers 21 and output drivers 20 begin driving data terminals 28 to a new data state (i.e., DATA 1 ). After another delay time sufficient to ensure that the new data state DATA 1 is stable, clock signal C50 returns low, shown at time t 3 of Figure 8. This again turns on transistor 72, reducing i BIAS to 50% of its maximum value, in this example, and thus reducing the DC current drawn through voltage reference and regulator 24. Adjustable bias current source Referring now to Figure 9, bias current source 26' according to an alternative embodiment of the invention will now be described in detail. Bias current source 26' provides for multiple levels of adjustment of the current i BIAS for voltage reference and regulator 24, controllable either by clock signals as in the case of dynamic bias circuit 60 described hereinabove, or by programming fuses. Bias current source 26' incorporates bias reference circuit 54 and current source transistor 52, connected to voltage reference and regulator 24 as before. In addition, as described hereinabove relative to Figure 7, transistors 72 and 74 are provided, to reduce the current i BIAS to 50% of its prior value when transistor 72 is turned on. In this case, however, the gate of transistor 72 is controlled by NAND function 73 which receives clock signal C50 at one input, and which receives the output of fuse circuit 75 on node FEN50* at another input. Fuse circuit 75 provides for the programmability of the state of transistor 72 in a permanent fashion. Such programmability may be useful in the early stages of the design and manufacture of memory 10, when the optimum, value of i BIAS has not yet been determined. In addition, programmability of the value of i BIAS is also desirable if the process variations in the manufacture of memory 10 vary widely enough that the optimum value of i BIAS is preferably set after initial test of the memory 10. For example, if memory 10 is processed to have very short channel widths, the value of i BIAS may be preferably reduced by programming fuse circuit 75 to maintain transistor 72 on at all times. Furthermore, one may program fuse circuit 75 to select a desired output slew rate. The construction of fuse circuit 75 may be accomplished in any one of a number of conventional ways. The example of Figure 9 simply has fuse 76 connected between V cc and the input of inverter 77, which drives node FEN50* from its output. Transistors 78 and 79 have their source/drain paths connected between the input of inverter 77 and ground. The gate of transistor 78 receives a power on reset signal POR, such that transistor 78 pulls the input of inverter 77 to ground upon power up of memory 10. The gate of transistor 78 is connected to the output of inverter 77 at node FEN50*. In operation, with fuse 76 intact, node FEN50* is held low by operation of inverter 77. With fuse 76 open, a pulse on line POR will pull the input of inverter 77 low, driving node FEN50* high, and turning on transistor 78 to maintain this condition. In operation, the output of NAND function 73 will be high if either clock signal C50 or node FEN50* is low. Accordingly, by not blowing fuse 76 open, node FEN50* will be held low, maintaining the output of NAND function 70 high and maintaining transistor 72 on unconditionally. With fuse 76 opened, clock signal C50 will control the state of transistor 72 as in the case of Figure 8 described hereinabove. Of course, it is contemplated that memory 10 may be implemented without clock signal C50, such that the state of transistor 72 is dependent solely upon the programmed state of fuse circuit 75. Bias current source 26' according to this alternative embodiment of the invention also includes transistors 72', 74' connected in series between node ISVR and ground, in similar fashion as transistors 72, 74 previously described. The gate of transistor 72 is similarly controlled by NAND function 73', responsive to the state of clock signal C67 and to fuse circuit 75' via node FEN67*. However, the size of transistor 74' is selected to be different from that of transistor 74 so that, when transistor 72' is turned on by either clock signal C67 or by fuse circuit 75', the current i BIAS is selected to be at a different fraction of its maximum value. For example, if the channel width of transistor 74' is one-half that of transistor 52 and of transistor 64 in bias reference circuit 54 (assuming the same channel length), then the effective channel width of the parallel combination of transistors 64, 74' will be 1.5 times the channel width of transistor 52. Accordingly, the value of i BIAS with transistor 74' turned on will be two-thirds that of its maximum value with transistor 74' turned off. Of course, other transistors of varying sizes may be similarly implemented into bias current source 26', if different values of current i BIAS are desired to be permanently programmed or clocked in at specific times of the memory cycle. In addition, for example, both of transistors 72, 72' may be simultaneously turned on to further reduce the current i BIAS . It is contemplated that other combinations of reduction in current will be apparent to those of ordinary skill in the art. According to this alternative embodiment of the invention, therefore, the value of the bias current i BIAS may be optimized for the particular design, for individual memory circuits depending upon the process parameters as determined by electrical test, or at specific points in time during the memory cycle. This optimization allows optimization of the tradeoff between maximum source and sink current and minimum output impedance for voltage regulator and reference 24, on the one hand, and the current drawn by voltage regulator and reference 24, on the other hand. In addition, the desired output slew rate may be selected in this optimization. Additional sink current path circuitry Referring now to Figures 12 and 13, alternative embodiments of the present invention, in which the sink current output impedance of output buffer bias circuit 22 is reduced still further, without increasing the DC power dissipation of the circuit, will now be described in detail. In each of the alternative embodiments of Figures 12 and 13, additional circuitry is provided that will conduct significant current when the load on line VOHREF requires output buffer bias circuit 22 to sink substantial current. As is apparent from the foregoing description, the DC power dissipation of voltage regulator and reference circuit 24 is primarily dictated by the value of the current i BIAS conducted by bias current source 26. The sink current capability of voltage regulator and reference circuit 24, is also limited to this bias current i BIAS , however. Accordingly, the circuit designer must tradeoff sink current capacity versus DC power dissipation. Each of the embodiment described hereinbelow relative to Figures 12 and 13, however, provide additional sink current capability without requiring an increase in the value of the current i BIAS , and thus without increasing the DC power dissipation of output buffer bias circuit 22. Referring first to Figure 12, output buffer bias circuit 22, constructed in the manner described hereinabove relative to Figure 3, is shown in combination with sink current path circuit 25 according to a first alternative embodiment of the invention. Sink current path circuit 25 includes p-channel sink transistor 100, which has its source connected to line VOHREF and its drain at ground. The gate of sink transistor 100 is controlled by the drains of p-channel transistor 102 and n-channel transistor 104; the source of transistor 102 is biased to V cc , while the source of transistor 104 is at ground. P-channel transistor 102 has its gate biased to an on condition, and as such acts as a load to transistor 104. In the example of Figure 12, the gate of p-channel transistor 102 is connected to the drain of current source transistor 52 (and thus to the sources of transistors 48, 50) in voltage regulator and reference circuit 24. As such, transistor 102 will be maintained in substantially an on-state. Alternatively, the gate of transistor 102 may be biased to ground, such that transistor 102 acts substantially as a linear load device to transistor 104. In this embodiment of the invention, the gate of transistor 104 is connected to the gate of transistors 44 and 46 in voltage regulator and reference circuit 24, receiving the voltage V 44 . Accordingly, transistor 104 turns on as the voltage V 44 rises. The operation of voltage regulator and reference circuit 24 discussed above relative to Figures 5 and 6 dealt with the switching of output terminal 28 from a low logic level to a high logic level, in which case voltage regulator and reference circuit 24 was required to source a significant current to output drivers 20 via output buffers 21; the worst case for this condition is, of course, when all output drivers 20 are switching in that direction. Sink current path circuit 25 of Figure 12 is directed to providing sink current to output buffers 21, which may occur when many or all output drivers 20 are to switch their respective output terminals 28 from a high logic level to a low logic level at high frequency (i.e., voltage regulator and reference circuit 24 being required to sink current at time when it is still sourcing current in a prior cycle). Sink current path circuit 25 will also serve to limit overshoot in the case discussed above relative to Figures 5 and 6, where the voltage on line VOHREF overshoots its desired level upon voltage regulator and reference circuit 24 sourcing current to the output buffers 21. When voltage regulator and reference circuit 24 is to sink current, the voltage V 44 at the gates of transistors 44, 46 (and, in this embodiment, transistor 104) will rise, since the sink current through transistor 50 increases virtually up to the current i BIAS . As the voltage V 44 increases, transistor 104 turns on more fully, which pulls the voltage at the gate of transistor 100 toward ground, turning on transistor 104. With transistor 104 turned on, an additional current path from voltage VOHREF to ground is provided, which will assist in pulling line VOHREF to its desired level from the overshoot condition in a timely fashion. The output impedance of voltage regulator and reference circuit 24 is thus reduced by sink current path circuit 25 according to this embodiment of the invention. Once the voltage on line VOHREF returns to its desired level, the voltage V 44 also returns to its steady-state value. In this condition, transistor 104 conducts less, such that the gate of transistor 100 is biased to a non-conductive state by load transistor 102. Referring now to Figure 13, sink current path circuit 25' according to an alternative embodiment of the invention will now be described in detail. The construction and operation of voltage regulator and reference circuit 24 in Figure 13 is similar to that described hereinabove. Sink current path circuit 25' according to this embodiment of the invention is based on a current mirror, and is of similar construction as voltage regulator and reference circuit 24. In this embodiment of the invention, sink current path circuit 25' includes p-channel transistor 110, which has its source connected to line VOHREF and its drain at ground, and thus which provides an alternate sink current path. The gate of transistor 110 is biased by n-channel transistors 112, 114, which have their source/drain paths connected in series between V cc and ground; the gate of transistor 110 is connected to the drain of transistor 114 and the source of transistor 112. The gate of transistor 114 is biased by line ISVR from bias reference circuit 54 described hereinabove. The gate of transistor 112 is biased by a current mirror circuit, as will now be described in detail. According to this embodiment of the invention, the current mirror of sink current path circuit 25' includes a reference leg consisting of p-channel transistor 144 and n-channel transistor 148 having their source/drain paths connected in series with one another; the source of transistor 144 is biased to V cc and the source of transistor 148 is connected to the drain of a current source transistor 152. The gate and drain of transistor 144 are connected together, while the gate of transistor 148 receives a divided voltage from the resistor divider 47, 49 of voltage reference and regulator 24 described hereinabove. On the mirror side, p-channel transistor 146 has its source biased to V CC and its gate connected to the gate and drain of transistor 144, while n-channel transistor 150 has its drain connected to the drain of transistor 146 and its source connected to the source of transistor 148 in the reference leg, which is connected to the drain of current source transistor 152. The drains of transistors 146 and 150 are connected to the gate of transistor 112, such that their drain voltage controls the conduction of transistor 112. The gate of transistor 150 is connected to line VOHREF. Current source transistor 152 has its source biased to ground, and receives the bias voltage ISVR at its gate, and conducts a controlled current i BIAS therethrough, similarly as transistor 52 in voltage reference and regulator 24 described hereinabove. Preferably, transistors 148 and 150 are balanced relative to one another, and transistors 144 and 146 are balanced relative to one another, such that the current mirror ratio is substantially unity. A non-amplifying current mirror ratio may be used for this current mirror, as significant source or sink current is not required therefrom. As a result, no offset voltage will develop. In operation, the current conducted by transistors 146 mirrors the reference current conducted by transistor 144, under the control of transistor 148, with the reference and mirrored currents summing to the current i BIAS conducted by transistor 152. The current conducted by transistor 150 is controlled according to the voltage on line VOHREF, however. Accordingly, if the voltage on line VOHREF rises, due to the switching of the output buffers 21 as described hereinabove, transistor 150 will be turned on more fully. This will tend to drop the voltage at the drains of transistors 146, 150 and at the gate of transistor 112, turning off transistor 112. Transistor 114, which is controlled by the bias voltage ISVR, will thus tend to pull the gate of transistor 110 closer to ground, turning on transistor 110 and thus providing an additional current sink path to line VOHREF. This will provide a reduced output impedance for voltage reference and regulator 24 in the sink current state, thus reducing the extent to which the voltage on line VOHREF will tend to overshoot its desired level. Variable output V OH control According to another alternative embodiment of the invention, selectability of the VOHREF limiting function is provided, either by way of a logic signal or by way of fuse programmability. According to this embodiment of the invention, it is contemplated that not all memories of the same design may be specified for use in combination with other integrated circuits using lower power supplies. For example, a subset of the memories may have a V OH maximum of 5.0 volts, while a different subset may have a V OH maximum limited to 3.3 volts. For purposes of manufacturing ease and inventory control, it is preferable to have a single integrated circuit design suitable for use as either, where the decision between 5.0 volt or 3.3 volt V OH maximum may be made at the latest possible stage of the manufacturing process. In addition, the suitability of specific memory chips for 3.3 volt operation may depend on process parameters, such as current drive, such that certain memories may not meet the 3.3 volt operating specification even if the VOHREF limiting function is enabled, but would meet the operating specification for memories with 5.0 volt V OH maximum. In this case, it would be desirable to have selectability of the VOHREF limiting function after electrical test. Further in the alternative, it may be useful to have a special test mode for memory 10, in which the VOHREF limiting function could be selectively enabled and disabled. Referring now to Figure 10, an alternative embodiment of the invention is illustrated in which voltage reference and regulator 124 is similarly constructed as voltage reference and regulator 24 described hereinabove, but may be disabled by way of an external signal, a special test mode signal, or programming of a fuse circuit. Those elements common to voltage reference and regulator 24 and voltage reference and regulator 124 are referred to by the same reference numeral, and will not be described again relative to voltage reference and regulator 124 of Figure 10. In addition to the previously described elements, voltage reference and regulator 124 includes p-channel transistors 82, 84, 89, and n-channel transistor 86, which force certain nodes to V cc or to ground in the event that the VOHREF limiting function is to be disabled, as indicated by the output of NOR gate 80 as will be described hereinbelow. Each of p-channel transistors 82, 84, 89 has its source biased to V cc , and its gate receiving line LIMOFF* from the output of NOR gate 80. The drain of transistor 82 is connected to the gates of transistors 44, 46 in the current mirror of voltage reference and regulator 124, the drain of transistor 84 is connected to line VOHREF at the output of voltage reference and regulator 124, and the drain of transistor 89 is connected to the input to bias reference circuit 54. N-channel transistor 86 has its drain connected to node ISVR in bias current source 26, has its source connected to ground, and has its gate receiving signal LIMOFF*, after inversion by inverter 85. According to this embodiment of the invention, pass gate 88 is provided between voltage PVBIAS and bias reference circuit 54, and is controlled by true and complement signals based on the signal LIMOFF*. In operation, if line LIMOFE* at the output of NOR function 80 is at a high logic level, transistors 82, 84, 86, 89 are all turned off and pass gate 88 is turned on; in this case, voltage reference and regulator 124 operates to limit the voltage at line VOHREF in the manner described hereinabove for voltage reference and regulator 24. However, if line LIMOFF* at the output of NOR function 80 is at a low logic level, transistors 82, 84, 86, 89 are all turned on and pass gate 88 is turned off. In this condition, line VOHREF is forced to 5.0 volts, and thus the drain voltage applied to output buffers 21 (and thus applied to the gate of pull-up transistors 32 in output drivers 20) is not limited to a reduced level. In order to minimize DC current drawn through voltage reference and regulator 124, certain nodes therein are also forced to particular voltages. In this example, the gates of transistors 44, 46 are pulled to V cc by transistor 82, thus turning off both of the reference and mirror legs in voltage reference and regulator 124. Pass gate 88 disconnects voltage PVBIAS from bias reference circuit 54, transistor 89 pulls the input to bias reference circuit 54 to V cc , and transistor 86 pulls node ISVR to ground, thus turning off transistors 52 and 58. Of course, the output of NOR function 80 may also be applied to nodes within offset compensating current source 28, bias reference circuit 54, and the like, as desirable. In this example of the invention, NOR function 80 receives three inputs, any one of which being at a high logic level will cause line LIMOFF* to be driven low. A first input is logic signal DIS, which may be generated elsewhere in memory 10, for example in timing and control circuitry 14; for example, a certain combination of inputs or instructions may be applied to memory 10 such that logic signal DIS is activated. A second input of NOR function 80, on node FDIS, is generated by fuse circuit 90. Fuse circuit 90 is constructed as described hereinabove relative to fuse circuit 75, such that node FDIS is at a low logic level with the fuse intact, and at a high logic level if the fuse is blown. According to this embodiment of the invention, a special test pad TP can also control the enabling and disabling of voltage reference and regulator 124 during electrical test in wafer form (i.e., prior to packaging). Test pad TP is connected to the input of inverter 91, which drives node TDIS received as an input of NOR function 80. Transistor 92 has its source/drain path connected between the input of inverter 91 and ground, and has its gate connected to node TDIS at the output of inverter 91. Transistor 93 has its source/drain path connected between the input of inverter 91 and ground, and its gate controlled by the power on reset signal POR. In operation, if test pad TP is held at V cc , inverter 91 will force node TDIS low. However, if test pad TP is left open or is connected to ground, upon power up transistor 93 will pull the input of inverter 91 low, forcing a high logic level on node TDIS which is maintained through operation of transistor 92. It is contemplated that test pad TP can thus control the enabling and disabling of voltage reference and regulator 124 during electrical test. Depending upon the result of such testing, test pad TP may be wire-bonded to V cc if voltage reference and regulator 124 is to be permanently enabled, or left open (preferably hard-wired to ground) if voltage reference and regulator 124 is to be permanently disabled for a particular memory 10. Such selective enabling and disabling of the V OH limiting function of the voltage reference and regulator according to the present invention is contemplated to greatly improve the manufacturing control of integrated circuits incorporating the function. In particular, integrated circuits corresponding to different specification limits may be manufactured from the same design, with selection of the maximum V OH voltage made late in the process, after electrical test. In addition, as noted above, fuse programming may be used to adjust the voltage divider presenting the input voltage to the voltage reference and regulator circuit, allowing additional tuning of the desired maximum V OH voltage. Embodiments of the invention can provide a voltage reference circuit that can source a relatively large current, while avoiding the development of an offset voltage therein. Embodiments of the invention can provide such a circuit having improved transient response while maintaining a stable output reference voltage. Embodiments of the invention can provide such a circuit which has improved switching performance. Embodiments of the invention can also provide such a circuit having a relatively low output impedance, such that modulation of the reference voltage is minimized when significant source and sink currents are demanded. Embodiments of the invention can further provided such a circuit in which the output impedance when a large sink current is demanded is not limited by the bias current source.

Claims: A voltage reference circuit, comprising: a first reference transistor (44), having a source/drain path connected on a first end to a power supply voltage, and having a gate connected to a drain end of its source/drain path; a second reference transistor (48) having a source/drain path connected between a second end of the source/drain path of the first reference transistor (44) and a common node, and having a gate for receiving a reference voltage; a first mirror transistor (46), having a source/drain path connected between the power supply voltage and an output node, and having a gate connected to the gate of the first reference transistor (44); a second mirror transistor (50), having a source/drain path connected between the output node and the common node, and having a gate; a bias current source (26), coupled to the common node, for conducting a current including the sum of the currents in the second reference transistor (48) and the second mirror transistor (50); a sink current path circuit (25,25'), connected to the output node, for conducting current responsive to the voltage at the output node exceeding a desired level, wherein the sink current path circuit (25,25') comprises: a sink transistor (100,110) having a conduction path connected between the output node and ground, and having a control electrode; a sink bias circuit (102,104,112,114) coupled to the output node and to the control electrode of the sink transistor ( 100,110), for turning on the sink transistor (100,110) responsive to a rise in the voltage at the output node. The voltage reference circuit of claim 1, wherein the sink bias circuit comprises: a load transistor (102,112), having a conduction path connected between a first voltage and the control electrode of the sink transistor (100,110), and having a control electrode biased to a bias voltage; and a sink control transistor (104, 144), having a conduction path connected in series with the conduction path of the load transistor (102,112), and having a control electrode coupled to respond to the voltage at the output node. The voltage reference circuit of claim 2, wherein the control electrode of the sink control transistor (104) is coupled to the gate of the first mirror transistor (46). The voltage reference circuit of claim 1, 2 or 3, wherein the first mirror transistor (46) and the first reference transistor (44) are p-channel field effect transistors. The voltage reference circuit of claim 4, wherein the load transistor (102) is a p-channel field-effect transistor and the sink control transistor (104) is an n-channel field effect transistor. The voltage reference circuit of claim 5, wherein the control electrode of the load transistor (102) is connected to the common node. The voltage reference circuit of claim 2, further comprising: a first sink reference transistor (144), having a source/drain path connected on a firs end to the power supply voltage, and having a gate connected to a drain end of its source/drain path; a second sink reference transistor (148) having a source/drain path connected between a second end of the source/drain path of the first sink reference transistor (144) and a sink common node, and having a gate for receiving the reference voltage; a first sink mirror transistor (146), having a source/drain path connected between the power supply voltage and the control electrode of the load transistor (112), and having a gate connected to the gate of the first sink reference transistor (144); a second sink mirror transistor (150), having a source/drain path connected between the control electrode of the load transistor (112), and the sink common node, and having a gate; and a sink bias current source (152), coupled to the sink common node, for conducting a current including the sum of the currents in the second sink reference transistor (148), and the second sink mirror transistor (150). The voltage reference circuit of claim 7, further comprising: a voltage divider (47,49), for generating the reference voltage applied to the gate of the second reference transistor (48) and to the gate of the second sink reference transistor (148). The voltage reference circuit of any of the preceding claims, wherein the second mirror transistor (50) has a current drive capability that is significantly greater than that of the second reference transistor (48). The voltage reference circuit of any of claims 1 to 9, further comprising an offset compensating current source (28), wherein the offset compensating current source (28) is connected to the second end of the source/drain path of the first reference transistor (44), for supplying a current thereto. The voltage reference circuit according to any one of the preceding claims, further comprising: a shift circuit (30) connected to bias the gate of the second mirror transistor (50) to ensure that the voltage at the output node is shifted upward. An output driver circuit for an integrated circuit, comprising: a pull-up drive transistor (32), having a conduction path coupled between a power supply voltage and an output terminal (28 j ,28 j-1 ...), and having a control electrode; circuitry for limiting the drive of the pull-up drive transistor responsive to a bias voltage (21); and a voltage reference circuit according to claim 8, or any of claims 9 to 11 when appended thereto, for producing the bias voltage, wherein the reference voltage generated by the voltage divider is based upon the power supply voltage, the voltage reference circuit further comprising a reference leg connected between the first reference transistor (44), the second reference transistor (48) and the common node, and a mirror leg connected between the first mirror transistor (46), the second mirror transistor (50) and the common node, said reference leg receiving the divided voltage from the voltage divider and conducting a reference current responsive thereto, said mirror leg conducting a mirrored current responsive to the reference current and deriving, at an output, the bias voltage based upon said mirrored current; wherein the bias current source is coupled to the reference leg and the mirror leg of the current mirror, for controlling the reference current and the mirror current conducted thereby; and wherein the sink current path circuit is coupled to the bias voltage output of the current mirror, for conducting sink current responsive to the bias voltage output exceeding a desired level. The output driver circuit of claim 12, wherein the circuitry for limiting the drive of the pull-up drive transistor (32) comprises: an output buffer (21 j ,21 j-1 ...), having an input for receiving a data signal, and having an output connected to the control electrode of the pull-up drive transistor (32), said output buffer connected to receive a bias voltage that is lower than the power supply voltage.

Inventors: MCCLURE DAVID C, MCCLURE, DAVID C.
Applicants: ST MICROELECTRONICS INC, STMICROELECTRONICS, INC.
Application year: 1995
Main classification: G05F1
IPC codes: G05F1, H03F3