Overload protection circuit for MOS power drivers

Abstract: The circuit (10) includes two regulating loops connected parallel to each other. The slow regulating loop (11) presents a lower first intervention threshold, and the fast regulating loop (12) a higher second intervention threshold. The slow regulating loop is low-gain and frequency-stable for accurately controlling the maximum value of the current (I D ) supplied by the driver (1) in the event of slow overloads or transient states. In the event of rapid overloads, the slow regulating loop fails to intervene effectively, current supply increases rapidly, and the fast regulating loop is turned on to rapidly discharge the parasitic capacitance of the driver (1). The resulting increase in current supply due to instability of the fast loop poses no hazard by virtue of the power transistor being saturated and so permitting effective intervention of the slow regulating loop (linear regulating region) which maintains the output current at the set nominal value.

Description: The present invention relates to an overload protection circuit for MOS power drivers, in particular NMOS and DMOS transistor high-side drivers, according to the preamble of claim 1. As is known, high-side power devices are interposed between the load and supply, and are formed by an N-channel MOS transistor (e.g. enhancement MOSFET or DMOS) with the drain terminal connected to the positive supply. One terminal of the load is connected to the source terminal of the transistor, and the other load terminal is grounded. Such an arrangement is shown in Figure 1 in which 1 indicates the power transistor, 2 the load, and D, G, S the drain, gate and source terminals respectively. To achieve maximum power at the load, the power transistor must operate in a linear region (like a resistor), i.e. the voltage V GS between the gate and source terminals must be higher than the voltage V DS between the drain and source terminals plus a threshold voltage V TH . That is: V GS > V DS + V TH or V G > V D + V TH V G > V CC + V TH where V G and V D respectively represent the gate and drain terminal voltages with respect to ground, and V CC is the supply voltage. In other words, the gate voltage V G must be higher than the supply voltage. To achieve the highest possible voltage at the load and hence a minimum voltage drop at the power transistor (and minimum power dissipation by the transistor), voltage V G must be considerably higher than supply voltage V CC . The gate voltage of currently used integrated circuits is normally required to be roughly 10 V higher than the supply voltage, which higher voltage is normally supplied by so-called booster or charge pump circuits. Numerous applications also require that the power outputs of the integrated circuit be provided with current limiting devices to safeguard the output transistor against overloading. A typical arrangement is shown in Figure 2 which shows a MOS drive transistor 1, load 2, and a regulating loop 3 comprising a current and/or voltage detecting element (sensing resistor 4) and a regulating element 5 with a threshold (represented in this case by a voltage source 6). The output of the regulating element is connected to the gate terminal of transistor 1. The dotted line in Figure 2 indicates the parasitic capacitance C GS between the source and gate regions of transistor 1. Regulating loop 3 controls current flow through drive transistor 1 and, when the current exceeds the predetermined threshold, begins drawing current I 1 to reduce voltage V GS of transistor 1 and so regulate the operating point and the current supplied by transistor 1. In the case of small MOS power transistors (with a current flow of a few tens of mA), the regulating element consists of a high-gain operational amplifier capable of responding rapidly to fast transient states at the load. In the case of medium or large MOS power transistors (with output currents ranging between a few tens of mA and several Amps), the regulating element cannot consist of a low-impedance-output operational amplifier, due to the high parasitic capacitance C GS typical of MOS transistors resulting in frequency instability of the regulating loop. Consequently, circuits operating with fairly low bias currents and limited gain are employed to ensure frequency stability of the regulating loop (capacitance C GS is used directly as a compensating capacitance, and is located at a high-impedance point). Nevertheless, the stability obtainable becomes increasingly critical as the load driven by the power transistor approximates a short-circuit. The load resistance in fact enters directly into the equation determining the dominant pole of the loop transfer function (loop gain) of the regulating circuit. As such, current I 1 for reducing voltage V GS in the event of an overload is limited (at most a few tens of µ A in the case of very large MOS transistors). In the case of a rapid overload at the output (e.g. a mechanical overload during low-inductance operation, typically with very short wiring close to the output pin of the integrated circuit, as in the case of a load supply wire shorting with a considerable mass of metal such as the bodywork of a vehicle), the output current increases extremely rapidly. In such a situation, voltage V GS must be reduced rapidly to bring the MOS transistor to the saturation region according to the statement: V GS < V DS + V TH that is: V G < V CC + V TH in which the transistor operates as a current source controlled by voltage V GS , i.e. I D = f(V GS ), where I D is the current through transistor 1. In fact, only alongside saturation operation of transistor 1 does the regulator operate as a linear regulator; and before the transistor reaches such a condition, the output current increases uncontrollably. To bring about the above condition, the parasitic capacitance C GS between the gate and source terminals must therefore be discharged from a value of roughly V CC + V TH + 10V to a value ranging between 0 V and V CC + V TH , depending on the value of the load. More specifically, in the case of an overload nearing ideal short-circuiting (very low resistance), the voltage at the gate terminal must be reduced by roughly V CC + 10V. Since, for a capacitor of capacitance C supplied with current I: I/C = dV/dT where dV is the voltage variation within time dT, in the present case this gives: I 1 /C GS = dV GS /dT and therefore dT = dV GS * C GS /I 1 For typical values of I 1 = 20 µ A, C GS = 400 pF and dV GS = 15 V, dT = 300 µ s. For larger transistors (higher C GS ), times in excess of a millisecond are obtained. This means that, within time interval dT, the output current may reach extremely high values limited solely by the resistance of the transistor and wiring, and normally resulting in damage to the driver. The standard practice for protecting power devices of the type in question is to turn off the power stage when the current supplied by it exceeds a given threshold. The driver remains off and can only be turned on again by external intervention at the control input. To this end, various solutions have been proposed based on measuring the voltage drop at the power transistor in addition to the output current, to determine the product and limit the total power dissipated by the transistor. Such solutions however cannot be applied to cases in which the load current is not permitted under any condition to fall below a guaranteed minimum (higher than the nominal current), i.e. in which the short-circuit current must correspond accurately with a specific value. To limit dissipation, in fact, the output current depends on the voltage of the transistor (as the transistor voltage increases, i.e. as the load resistance decreases, the output current is reduced to maintain the voltage-current product constant). Moreover, in the case of MOS transistors of considerable size, the above known solutions do not always respond adequately to extremely rapid transient states (rapid short-circuiting) at the load. US-A-4 809 122 teaches a protection device comprising three protection loops: a first protection subcircuits 28, fast acting, which is a regulator and has the task of regulating the current flowing through the MOS transistor at initial turn-on; a second protection subcircuit 36 which is a switching-type stage and has the task of turning off the MOS transistor when the voltage is in excess of the normal full load saturation voltage; and a third protection subcircuit 82 which is also of the switching type and has the task of sensing long term increased current flow and of turning off the MOS transistor when the normal steady state current reaches a preset high. Thus, if an excess current condition is detected during normal operation, the known circuit provides for turning the MOS transistor off, which, in some applications, is to be avoided. It is an object of the present invention to provide a protection circuit capable of responding rapidly, even in the event of rapid overloading, without turning off the power stage. According to the present invention, there is provided an overload protection circuit for MOS power drivers, as claimed in Claim 1. In practice, according to the present invention, the slow regulating loop, being low-gain, is frequency-stable and accurately controls the maximum current supplied by the driver in the presence of slow overloads or transient states. Conversely, in the presence of rapid overloads which cannot be catered to effectively by the slow regulating loop, the fast regulating loop is activated to rapidly discharge the parasitic capacitance of the power transistor. The fast loop results in oscillation of the system but nevertheless brings the power transistor to the saturated condition (linear regulating region) in which the slow regulating loop effectively maintains the output current at the predetermined rated value. A preferred non-limiting embodiment of the present invention will be described by way of example with reference to the accompanying drawings, in which: Figure 1 shows a known power driver to which the present invention is applied; Figure 2 shows a known protection circuit; Figure 3 shows a diagram of the protection circuit according to the present invention; Figure 4 shows the electric diagram of one embodiment of the protection circuit according to the present invention. In Figure 3, the protection circuit according to the present invention is indicated as a whole by 10, and comprises two parallel current regulating loops 11 and 12. More specifically, the first (slow) regulating loop 11 provides for ensuring accuracy of the current supply in the. event of a short-circuit or overload; presents a low gain; and, as in known solutions, comprises a regulating element 15, the inputs of which are connected to the two terminals of a sensing resistor 4 series connected between V CC supply line 17 and power transistor 1. The intervention threshold of regulating element 15 is represented by a voltage source 16, and the output of regulating element 15 (in the form of an OTA - Operational Transconductance Amplifier - i.e. an operational amplifier with an output current I 1 ) is connected to the gate terminal of power transistor 1. The second (fast) regulating loop 12 presents a very high gain, and hence an output current I 2 much higher than I 1 , and comprises a a regulating element 20 formed substantially by a comparator, the inputs of which are connected to the two terminals of sensing resistor 4, and the output of which is connected to the gate terminal of power transistor 1. The noninverting input of comparator 20 is connected to the high terminal (connected to line 17) of sensing resistor 4 via a voltage source 21 representing the intervention threshold of regulating loop 12, which is selected higher than the intervention threshold of regulating loop 11 according to the statement: V REF1 < V REF2 where V REF1 and V REF2 are the voltages existing at the non-inverting inputs of elements 15 and 20. Being high-gain, the second regulating loop provides for ensuring correct operation of protection circuit 10 as explained below. In the event of a not particularly rapid short-circuit or overload (slow transient state), only first regulating loop 11 is operated, due to its lower intervention threshold. That is, upon the current through sensing resistor 4 exceeding a predetermined value corresponding to voltage V REF1 , regulating element 15 is operated, which, by means of current I 1 , reduces the voltage at the gate terminal of power transistor 1, and regulates the operating point of the transistor to prevent an excessive increase in current I D . Being a slow transient state, first regulating loop 11 operates correctly, no oscillation is produced, and the voltage at the terminals of sensing resistor 4 at no time exceeds the intervention threshold of second regulating loop 12. Conversely, in the event of a rapid overload at the output of power transistor 1 (terminal S), output current I D rapidly exceeds the threshold v REF1 of the first regulating loop, which fails to operate effectively, and subsequently also exceeds the second threshold (intervention threshold V REF2 of second regulating loop 12). Consequently, second loop 12 is activated, which, presenting a very high gain (high output current I 2 ), rapidly discharges the parasitic capacitance C GS of power transistor 1 (typically in a few µ s), thus rapidly reducing the output current I D of the transistor and so preventing it from being damaged. Discharge of the parasitic capacitance C GS of power transistor 1 produces an oscillation in turn resulting in a further increase in the output current I D of the transistor. This increase is detected by first regulating loop 11 which, this time, is in a betted condition to operate by virtue of power transistor 1 presenting a discharged gate region and being in the saturation region in which it operates as a current source controlled by voltage V GS , In this condition, first regulating loop 11 therefore operates as a linear regulator. Moreover, as the current, at this point in its increase, is still low, power transistor 1 is in no danger of being damaged despite the slow nature of first regulating loop 11. In this second phase, first regulating loop 11 therefore operates properly, preventing an excessive increase in the output current I D of the transistor, and accurately ensuring the regulated current value as well as its own frequency stability, while second regulating loop 12, being below its intervention threshold, remains inactive. In other words, the faster, second regulating loop 12 operates solely in the event of rapid, high transient states in output current I D , by rapidly discharging the gate capacitance C GS of power transistor 1 and bringing the system into a condition better suited to ensure accurate operation of the first regulating loop. Figure 4 shows an example embodiment of protection circuit 10 according to the present invention. As shown in Figure 4, the gate terminal of power transistor 1 is connected via transistor 30 to a charge pump circuit 31 formed in any known manner and therefore not shown in detail. Power transistor 1, transistor 30 and charge pump circuit 31 (possibly with control elements not shown) together form a driver 29 having an output 28 connected to a load (not shown) supplied with current I D . Transistor 30 is a split-collector PNP transistor with the emitter terminal connected to the output of charge pump 31, the base terminal connected to the gate terminal of power transistor 1, a first collector terminal connected to its own base terminal, and the second collector terminal connected to a current mirror circuit 32 forming part of first regulating loop 11. First regulating loop 11 substantially comprises a differential stage 33 comprising a first and second NPN type transistor 34, 35, and a third and fourth split-collector PNP type transistor 36, 37. More specifically, transistor 34 presents the collector terminal connected to supply line 17, the base terminal connected to the drain terminal of power transistor 1, and the emitter terminal connected to the emitter terminal of transistor 36. Transistor 35 presents the collector terminal connected to supply line 17, the base terminal connected to a node 40, and the emitter terminal connected to the emitter terminal of transistor 37. Node 40 is set to a fixed voltage representing the threshold voltage of first regulating loop 11, and is connected to one terminal of a resistor 41 (the other terminal of which is connected to supply line 17) and to a current source 42 which therefore determines the voltage V R at the base terminal of transistor 35. Transistors 36 and 37 present the base terminals connected to each other at node 43, a first collector terminal short-circuit-connected to the respective base terminal, and a second collector terminal connected to a respective node 45, 46. A bias current source 44 is interposed between node 43 and ground. Node 45 is connected to one terminal of a resistor 47 (the other terminal of which is grounded) and to the emitter terminal of an NPN type transistor 48 forming part of current mirror circuit 32 and presenting the collector terminal connected to the gate terminal of power transistor 1. Node 46 is connected to one terminal of a resistor 49 (the other terminal of which is grounded) and to the emitter terminal of a diode-connected NPN type transistor 50 also forming part of current mirror circuit 32 and presenting the collector terminal connected to the second collector of transistor 30. Second regulating loop 12 and first regulating loop 11 share the input stage (formed by transistors 34, 35) of differential stage 33, as well as reference voltage source 41, 42. In particular, second loop 12 comprises a first pair of PNP type transistors 55, 56 and a second pair of NPN type transistors 57, 58 forming a current mirror circuit 59. More specifically, transistor 55 presents the emitter terminal connected to the emitter terminal of transistor 34, the base terminal connected to node 43, and the collector terminal connected to the collector terminal of transistor 57. Transistor 56 presents the emitter terminal connected to the emitter terminal of transistor 35, the base terminal connected to node 43, and the collector terminal connected to the collector terminal of transistor 58. Transistor 57 of current mirror circuit 59 is diode-connected, and presents a grounded emitter terminal and the base terminal connected to the base terminal of transistor 58 which also presents a grounded emitter terminal. Transistors 57 and 58 present an area ratio of 2, so that circuit 59 represents a mirror with a 1:2 ratio, i.e. transistor 58 tends to draw twice the current of transistor 57, to permit regulating loop 12 to operate at a higher threshold than loop 11 despite sharing reference voltage V R , as described in more detail below. The collector terminal of transistor 58, defining node 60, is connected to the gate terminal of a DMOS transistor 62 having a grounded source terminal and the drain terminal connected to the gate terminal of power transistor 1. The Figure 4 circuit operates as follows. Under normal operating conditions of driver 29, i.e. with a low voltage drop at the terminals of sensing resistor 4, the potential at the base terminal of transistor 34 is greater than the potential V R at the base terminal of transistor 35, so that transistors 37 and 56 are off, and all the current of bias source 44 is supplied to transistors 36, 55. In known manner, resistor 47 receives the current supplied by transistor 36, and its voltage drop is greater than that (zero) at the terminals of transistor 50, so that transistor 48 is off, current I 1 is zero, and the first (slow) regulating loop is off. Similarly, transistor 56 being off, transistor 58 of mirror 59 and DMOS transistor 62 are also off, so that I 2 is also zero and the second regulating loop is also off. As the output current I D of power transistor 1, and hence the voltage drop at sensing resistor 4, increases, and if the increase is slow, the voltage drop between the base terminals of transistors 34 and 35, in known manner, brings the differential stage to a more balanced position, or even to a position of unbalance towards transistor 35, thus gradually turning on transistor 35, gradually reducing the voltage at resistor 47, and gradually increasing the voltage at resistor 49. When the voltage at resistor 46 exceeds that of resistor 47, transistor 48 comes on and begins conducting current I 1 , the value of which depends on the unbalance of differential stage 33, so that slow loop 11 accurately regulates output current I D . At this phase, fast regulating loop 12 is inactive. In fact, in view of the area ratio of transistors 57, 58, transistor 58 remains off until the input stage of differential stage 33 is so unbalanced that the current supplied by transistor 56 is twice that through transistor 55, which unbalance, in the event of a slow transient state of the overload, is prevented by operation of slow regulating loop 11. As already seen, in the event of a rapid overload at output 28 of driver 29, slow regulating loop 11, even after being turned on, is incapable of preventing a further rapid increase in current I D , which increase therefore results in equally rapid unbalance of the input differential stage and an increase in the current through transistors 56 and 58. Upon the current supplied by transistor 56 exceeding twice the current through transistor 55 (balance point of the fast regulating loop differential stage), the excess current supplied by transistor 56 is supplied to DMOS transistor 62, the gate capacitance of which charges, and DMOS transistor 62 in turn begins conducting and so drawing current I 2 . In view of the high gain of fast loop 12, said current is high and rapidly discharges the gate region of power transistor 1 to eliminate current I D . As already stated, an oscillation is produced (inversion of the direction of current I 2 and a further increase in current I D ) which nevertheless saturates power transistor 1, thus assisting operation of the slow regulating loop which therefore blocks the oscillation and provides for accurately regulating current I D . The advantages of the circuit described are as follows. In particular, it provides for short-circuit and overload protection without turning off the power stage, so that, upon elimination or termination of the overload condition, the driver continues operating with no external intervention at the inputs required for resetting the circuit. Following fast intervention of loop 12, the slow regulating loop 11 operates to a high degree of precision, so that, after the very short first phase, the current supplied in the event of an overload is predictable. The Figure 4 embodiment provides for reducing the overall size of the circuit by optimizing the layout and sharing a number of elements between the two regulating loops. Moreover, as the two regulating loops share the voltage source defining the intervention threshold of the first loop, and as the threshold of the second loop depends on the emitter area ratio of transistors 57 and 58, the threshold ratio of the two regulating loops is fixed and therefore independent of variations in temperature, supply voltage, reference threshold voltage V R , process spread, or any other interference equally involving the two loops. The arrangement shown relative to the second regulating loop also provides for extremely rapid intervention: the differential stage defining the intervention threshold drives a DMOS transistor acting directly on the gate of the power device and discharging its low-impedance capacitance. Clearly, changes may be made to the circuit as described and illustrated herein without, however, departing from the scope of the present invention. In particular, the circuit according to the present invention may also be applied to power transistors other than those described, or to low-side drivers; and the second regulating loop may be implemented in any appropriate manner.

Claims: An overload protection circuit (10) for a power MOS transistor driver (1) having a control terminal (G) at a control potential and supplying a drive current (I D ) to a load, said circuit (10) comprising: a slow, low-gain regulating loop (11) comprising a first operating element (15; 34-37) operative above a first predetermined voltage threshold corresponding to a first value of the drive current (I D ), the slow, low-gain regulating loop (11) including a measuring element (4) for measuring variations of the drive current (I D ) and for generating a current limiting signal (I 1 ) for regulating said control potential without turning off said power MOS transistor driver (1); a fast, high-gain regulating loop (12) connected in parallel with said slow, low-gain regulating loop (11) and comprising a second operating element (20; 34, 35, 55-58) operative above a second predetermined voltage threshold higher than said first predetermined voltage threshold and corresponding to a second value of the drive current (I D ) higher than said first value, the fast, high-gain regulating loop (12) detecting rapid variations in said drive current (I D ) and generating a rapid control signal (I 2 ) for rapidly discharging the capacitive gate region of the power MOS transistor driver (1) and rapidly reducing said control potential without turning off said power MOS transistor driver (1). A circuit according to claim 1, characterized in that said fast, high-gain regulating loop (12) comprises a current path between said control terminal (G) and a reference potential line; and a power switch (62) located along said current path, for generating a high discharge current (I 2 ) for said MOS power device (1). A circuit according to claim 1 or 2, characterized in that said first and second regulating elements of said slow, low-gain and fast, high-gain regulating loops (11, 12) each comprise a differential circuit (15, 20) having a first input connected to said measuring element (4), a second input connected to a respective threshold circuit (16, 21) and an output connected to said control terminal (G) of said MOS power device (1). A circuit according to claim 3, characterized in that said differential circuits (15, 20) of said slow, low-gain and fast, high-gain regulating loops (11, 12) have a common input stage (34, 35, 44) connected to a common threshold voltage source (40-42); and said differential circuit (34, 35, 55-58) of said fast regulating loop has a balance point at a higher value of said drive current (I D ) with respect to said differential circuit (33) of said slow regulating loop. A circuit according to claim 4, characterized in that said differential circuit (34, 35, 55-58) of said fast, high-gain loop (12) has a first and second branch (55; 56) connected to said common input stage (34, 35, 44) and to a current mirror circuit (59) with a predetermined area ratio; said current mirror circuit having an output (60) connected to a power switch (62) interposed between said control terminal (G) of said MOS power MOS driver (1) and a reference potential line.

Inventors: COLANDREA FRANCESCO, POLETTO VANNI, COLANDREA, FRANCESCO, POLETTO, VANNI
Applicants: ST MICROELECTRONICS SRL, STMICROELECTRONICS S.R.L.
Application year: 1994
Main classification: H03K17
IPC codes: H03K17