A speech amplifier associated with a LED

Abstract: The invention relates to a transmit amplifier for a telephone set without self supply, comprising a LED (1) having an anode connected to a first positive terminal (A) of the telephone line and means for determining the ratio between the line current (IL) and the line voltage (VL). The LED (1) is supplied with a constant current and this current is derived when the voltage (VL) across the terminals (A, B) of the line becomes lower than a predetermined value (Vs1).

Description: The invention relates to a speech transmit amplifier provided with a light emitting diode (LED). It more specifically applies to the control of the LED inside such an integrated amplifier in a telephone set. In a telephone set, an electroluminescent diode is generally used for lighting the key-pad, or for indicating a hold or dialling condition. Commonly, the telephone set does not comprise its own power supply but is fed by the current in the telephone line, this line current being also used for feeding the LED. The telephone signal comprises a DC component, provided by the telephone exchange and sent on the telephone line to be used by the subscribers' telephone sets. The DC component of the supply signal is variable as a function of the line length or other line parameters. The signal also comprises an AC component due to the modulation of the speech signal. Conventionally, a predetermined fraction of the line current is used, inside the amplifier, to supply the LED. This fraction is usually about 25% of the line current. As the line current varies as a function of the line characteristics, the LED illumination is not constant. Indeed, this illumination is low for long lines and very low for parallel connections. On the contrary, if the line is short, the illumination is too strong and reduces the lifetime of the LED. Additionally, as the current oscillates about the DC component, the use of a fixed portion of the line current to control the LED causes the illumination to vary for a given line. Furthermore, the presence of the LED causes the transmitted signal to be distorted for high negative swings of the speech signal. In some cases, a very small but fixed part of the line current is also used to light up the LED. But this current cannot be large, otherwise it will upset the voltage-current characteristics of the line. Therefore, in such cases, the LED illumination is poor. US-A-4 564 729 discloses a speech network and an illumination circuit for a telephone set without self supply which comprises a LED having an anode connected to a first positive terminal of the telephone line and means for varying the current in the LED with a transistor in series with the LED. It is an object of the invention to palliate the drawbacks of the existing amplifiers as regards the LED control. Another object of the invention is to provide a voice amplifier that feeds a LED with a constant current. Another object of the invention is to avoid the distortion of the speech signal caused by the presence of a LED in the voice amplifier. To attain these objects, the invention provides for a transmit amplifier for a telephone set without self supply, comprising a LED having an anode connected to a first positive terminal and the telephone line and means for determining the ratio between the line current and the line voltage, comprising supply means for supplying LED with a constant current and deriving means comprising a transconductance amplifier whose feedback loop controls a switch that derives the current flowing in the LED when the voltage across the terminals of the line becomes lower than a predetermined value. According to an embodiment of the invention, the supply means comprise a first operational amplifier measuring the current in the LED and whose feed-back loop limits this current to a value determined by a reference current generator. According to an embodiment of the invention, the cathode of the LED is connected to the non-inverting input of the first amplifier. According to an embodiment of the invention, the reference current generator comprises a bipolar PNP transistor having an emitter connected to the first terminal of the line and a collector connected to the inverting input of said first amplifier. According to an embodiment of the invention, the deriving means comprise a transconductance amplifier whose feed-back loop controls a switch that derives the current of the LED. According to an embodiment of the invention, the switch comprises an NPN transistor whose collector is connected to the first terminal of the line and whose emitter is connected, through a low value resistor, to the cathode of the LED. According to an embodiment of the invention, the deriving means also disconnect said supply means. According to an embodiment of the invention, the means for determining the ratio between the line current and the line voltage comprise a second operational amplifier whose inverting input receives the speech signal to be transmitted, added to a DC component of the line voltage provided by a telephone exchange. According to an embodiment of the invention, the feed-back loop of the second amplifier comprises two Darlington-connected NPN transistors and two diodes, the collector of the second transistor being connected to the inverting input of the transconductance amplifier and the non-inverting input of the transconductance amplifier being connected to the anode of the second diode. As the LED according to the invention is supplied with a constant current, the telephone sets of various subscribers will have the same characteristics whatever be the line length. As the LED current is shunted when the voltage on the telephone line becomes too low, the presence of the LED in the speech transmit amplifier does not distort the AC signals. Thus, one obtains a speech transmit amplifier whose operation is not modified by the presence of a LED, while optimizing the operation of this LED. Those objects, features and advantages, and others of the invention will be explained in detail in the following description of preferred embodiments made non-limitatively in connection with the attached drawings wherein: figure 1 schematically shows an embodiment of a speech transmit amplifier according to the invention; figure 2 illustrates the current voltage curve of the transmit amplifier according to the invention; figure 3 illustrates the current in the LED as a function of the line current; figures 4-7 are chronograms illustrating the potentials across the LED for various amplitudes of the speech signal; and figure 8 illustrates a detailed implementation of the circuit of figure 1. The speech transmit amplifier shown in figure 1 mainly comprises three feed-back loops. The first loop controls the ratio between the line current IL and the line voltage VL. The second loop measures the current I LED flowing in the LED 1 and limits same at a predetermined value. The third loop authorizes a suitable swing of the speech signal during the negative alternations, by deriving the current flowing in the LED 1. The telephone line is symbolized by a generator 2 between two terminals A and B of the circuit, terminal B corresponding to the circuit ground. The first loop comprises a first operational amplifier A1 3. The non-inverting input of amplifier 3 is connected to a first positive terminal A of the circuit through a voltage generator 4 and a resistor R1. The voltage generator 4 corresponds to two input terminals M+ and M- in the circuit of the amplified signal from the telephone microphone (not shown). The negative input terminal M- of the microphone signal is connected to the non-inverting input of amplifier 3 and the positive input terminal M+ to resistor R1. The inverting input of amplifier 3 is connected to terminal A through a first polarization current generator 5. This inverting input is also connected to the anode of a diode D1 serially connected with a diode D2. The cathode of diode D2 is connected to the ground terminal B of the circuit through a resistor R5 and to the positive input terminal M+ of the microphone signal 4 through a resistor R2. The feed-back loop of the operational amplifier 3 comprises the two diodes D1 and D2 and two NPN transistors T1, T2. The output of amplifier 3 is connected to the base of transistor T1, the collector of which is connected to terminal A. The emitter of transistor T1 is connected to the base of transistor T2 whose collector is connected to terminal A through a resistor R3 and LED 1. The emitter of transistor T2 is connected to the cathode of diode D2. A capacitor C2 for frequency compensation of the first loop is connected between the collector of transistor T1 and the base of transistor T2. The second loop comprises a second operational amplifier A2 6. The non-inverting input of amplifier 6 is connected to the cathode of LED 1. Its inverting input is connected to terminal A through a reference current generator 7 and to the collector of transistor T2 of the first loop through a resistor R4. The feed-back loop of amplifier 6 comprises the resistor R4, a diode D3 and three bipolar transistors T3, T4, T5. The output of amplifier 6 is connected to the base of a PNP transistor T5 whose collector is connected to terminal B. The emitter of transistor T5 is connected to the cathode of diode D3, the anode of which is connected to the base of the NPN transistor T4. The base of transistor T4 is also connected to terminal A through a second polarization current generator 8. The collector of transistor T4 is connected to terminal A and its emitter is connected to the base of the NPN transistor T3 whose collector is also connected to terminal A. The emitter of transistor T3 is connected to the collector of transistor T2 of the first loop and, through resistor R4, to the inverting input of amplifier 6. A capacitor C3 for frequency compensation of the second loop is connected between the base of transistor T5 and terminal A. The third loop comprises a transconductance amplifier gm 9. The non-inverting input of amplifier 9 is connected to the anode of diode D2 of the first loop. The inverting input of amplifier 9 is connected to the collector of transistor T2 of the first loop. The feed-back loop of the transconductance amplifier 9 comprises transistor T3 of the second loop and a PNP transistor T6. The output of amplifier 9 is connected to the base of transistor T6 whose emitter is connected to terminal A. The collector of transistor T6 is connected to the base of transistor T3, the emitter of transistor T3 being connected to the collector of transistor T2 and to the inverting input of amplifier 9. A resistor R6 serially connected with a capacitor C4 is connected between the output of amplifier 9 and its biasing input, so as to frequency compensate the third loop. The circuit also comprises a positive temperature coefficient current source 10 between terminals B and M+. For its dynamic operation, the circuit also comprises capacitors. A capacitor C1 is parallel-connected with the current generator 10, its negative terminal being connected to terminal B. The capacitor C1 provides AC ground at terminal M+. A capacitor C5 serially connected with a resistor R7 is connected between terminals A and B. A capacitor C6 is connected between terminals A and B. Capacitors C5, C6 and resistor R7 model the AC termination of the exchange. Figure 2 shows the voltage-current characteristic curve between terminals A and B of the circuit. This characteristic is determined by the first loop, that is by the current source 10, the operational amplifier 3, resistors R1, R2, R5, diodes D1, D2 and transistors T1, T2. The line voltage VL is equal to the sum of the voltages across terminals R1, R2, R5. The voltage across R5 is equal to the product of the line current IL by resistance R5. In the absence of speech signal between terminals M+ and M-, the voltage across resistor R2 is equal to 2Vbe where Vbe represents the base-emitter voltage drop of a transistor (that is the voltage drop across the forward biased diodes D1 and D2). The voltage across the resistor R1 is equal to the product of resistor R1 by the sum of the current Idc provided by the source 10 and the current circulating in resistor R2. This current in resistor R2 is equal to 2Vbe/R2. Therefore, the relation between the line voltage VL and the line current IL can be written: VL0 = IL*R5 + 2(1+R1/R2)Vbe + Idc*R1 This characteristic curve presents, from a value V0 corresponding to 2(1+R1/R2)Vbe + Idc*R1, a slope determined by resistance R5. The current source 10 compensates the variations of the value of the base-emitter voltages Vbe as a function of the operating temperature. So V0 is a temperature compensated reference voltage. At low currents, i.e. IL<IS1, V0 is reduced by reducing Idc gradually. This is controlled by an external circuit. In dynamic operation, that is in the presence of a speech signal, the voltage of line VL oscillates around a value VL0 determined by the above relation. The current control of LED 1 is made through the second loop, that is the current source 7, operational amplifier 6, resistors R3, R4, diode D3 and transistors T3, T4, T5. Resistor R3 has a low value and is used for measuring the current I LED in LED 1 (flowing in the components 1, R3, T2 and R5). This current is compared with a reference current generated by the generator 7 which flows in resistor R4. This second feed-back loop aims at having a limited portion of the line current IL to flow in the LED 1. So, as shown in figure 3, as long as the line current IL is lower than a threshold value Is2, the current in LED 1 varies like the current IL. As soon as the line current IL is sufficient, the current in LED 1 is limited to this threshold value Is2. The threshold value Is2 is determined by the values of resistors R3 and R4 and of the current Iref provided by the reference current source 7: Is2 = Iref*R4/R3 The remaining portion of the line current, that is the difference between the line current IL and the threshold current Is2, flows in transistor T3. However, if IL becomes lower than Is2, the loop saturates. Therefore, transistor T3 cuts off and the input voltages of the operational amplifier 6 are no longer equal. Then, the current in LED 1 is equal to IL. The third loop aims at avoiding a distortion of the transmitted signal for high amplitude swings of the speech signal. It derives the current of the LED 1 when the negative swings of the speech signals have a value such that the line voltage VL is lower than a threshold voltage Vs1. Accordingly, the transistor T2 does not saturate and does not cause a distortion of the transmitted signal. The two inputs of the transconductance amplifier 9 are respectively connected to the collector and, through diode D2, to the emitter of transistor T2. A minimum voltage drop, equivalent to Vbe, is accordingly permanently maintained between the collector and the emitter of transistor T2 if the line voltage gets lower than the threshold value Vs1. Indeed, in this case, the feed-back loop of amplifier 9 causes transistor T6 to be on. Thus, transistor T3 shorts the serial connection of LED 1 and resistor R3. This third loop also cancels, through transistor T4, the operation of the second loop that controls LED 1. When the voltage VL is higher than the threshold voltage Vs1, the third loop saturates. Transistor T6 cuts off. Thus, this third loop is not active during the positive swings or the low value negative swings of the line voltage VL. The effect of the third loop is particularly useful when the reference voltage V0 is low. Under such conditions, the risk of saturating transistor T2 during the negative swings increases if LED 1 is not shunted since V0 is also equal to the LED drop plus the collector-emitter drop of the transistor T2 plus the drop in the resistor R3 as forced by the first loop. Figure 3 shows the current Iled in LED 1 as a function of the line current IL in the absence of speech signals. As long as the line current IL does not reach a value Is1 corresponding to the threshold voltage Vs1, the LED 1 is shunted (phase 1) and no current flows therein. This results from the action of the third loop, the second loop being saturated. When the line current is comprised between the threshold current Is1 and the threshold current Is2, Iled-IL (phase 2). This is the action of the only first loop, the two other loops not being operative. When the line current IL is higher than the threshold value Is2, Iled is limited to Is2 (phase 3). This is the action of the second loop, the third loop being saturated. Figures 4-7 are time diagrams illustrating the operation phases as a function of the speech signal swings. These figures represent the line voltage VL that corresponds to the anode potential of LED 1 and the potential Vk on the cathode of LED 1. In these figures, the line current IL is assumed to be higher than the threshold value Is2 in the absence of speech signal. In other words, we are in the low slope portion of the voltage-current characteristic of the line. Figure 4 shows small swings of the line voltage VL, that is a speech signal having a low amplitude around VL0. In this case, the LED 1 is permanently conductive and Vk=VL-VD, VD being the nominal voltage drop of the LED 1. We are in the phase 3 of figure 3 for which the second loop is operative while the third loop is not operative. Figure 5 illustrates higher swings of the speech signal around the value VL0 such that the negative swings of VL reach the threshold value Vs1. In this case, the third loop gets operative and shunts the LED 1 (phase 1). The collector-emitter voltage of transistor T2 is maintained at Vbe by the third loop. Accordingly, the potential Vk of the cathode of LED 1 is also maintained at a voltage VS2 which is equal to Vbe plus the voltage drop in resistor R5 (VS2-Vbe+ILR5). The drop across the LED, VL-Vk, is now less than its normal value VD. This is due to partial shunting of the LED. This situation continues for still higher signal excursions till the LED is fully shunted, i.e. the LED drop is equal to Vbe(3)+VCEsat(6), where Vbe(3) is the base-emitter voltage of the transistor T3 and VCEsat(6) is the saturation collector-emitter voltage of the transistor T6. If the negative swings of the speech signal cause the line voltage VL to be at a still lower value, the voltage Vk of the cathode of LED 1 cannot be maintained at VS2. This is because the third loop now saturates with the LED fully shunted and therefore, the collector-emitter voltage of transistor T2 drops below Vbe. The cathode voltage of LED 1 now goes down to VL-Vbe(3)-VCEsat(6). As long as the transistor T2 does not saturate, Vk follows the variation of the line voltage VL, this explains the clip in Vk at the bottom as shown in figure 6. We are always in phase 1 wherein the first and third loops are active, providing a transmission without distortion of the speech signal. Figure 7 shows speech signal oscillations that are such that the line voltage VL gets lower even further and causes the saturation of transistor T2. The speech signal can no longer be transmitted without distortion even in the absence of LED 1 which is anyway fully shunted. The value ILR5 + VCEsat(2) (where VCEsat(2) is the saturation collector-emitter voltage of the transistor T2) for voltage VL at its negative excursion thus constitutes the limit of the operation of the amplifier according to the invention to transmit an undistorted signal. As an example, the LED 1 may have a nominal voltage VD equal to 2 V. We assume that there is on the line a DC line voltage VL0 varying between 3.5 and 5.5 V for IL - 20 mA and 100 mA respectively, and a speech signal whose swings are not higher than 2 V peak. Then, it is possible to select for the resistors and capacitors the following values: R1 : 17 kΩ; C1 : 4.7 µF; R2 : 39 kΩ; C2 : 5 pF; R3 : 6 Ω; C3 : 20 pF; R4 : 5 kΩ; C4 : 20 pF; R5 : 20 Ω; C5 : 100 µF; R6 : 9 kΩ; C6 : 10 nF; R7 : 600 Ω. For IL = 20 mA, the collector-emitter voltage of transistor T2 is about 1.1 V. It is equal to 1.5 V when IL - 100 mA. In the absence of the third loop, the voice signal would be distorted as soon as the speech signal swings attain a peak value of about 0.9 V (1.1 V - VCEsat(2) at IL - 20 mA and 1.3 V at IL = 100 mA). With the invention, this signal is not distorted. This is due to the fact that LED drop of 2 V is shunted with a lower voltage equal to 0.9 V (Vbe(3) + VCEsat(6)). Therefore, there is at least 2.0 - 0.9 = 1.1 V more room for the transistor T2 to saturate. Figure 8 shows an example of a detailed implementation of the circuit of figure 2. This circuit comprises the terminals A, B, M+ and M-. This circuit also comprises a terminal Vcc which gives a separate and stable supply for the amplifier A1 3. The Vcc supply is obtained from the line by means of a resistor R12 and a capacitor C7 serial connected between terminals A and B. The Vcc supply is also used to generate the bias currents of some transistors of the circuit and especially those constituting the amplifier A1 3. The details of the circuitry for obtaining those currents are not shown, only two terminals, respectively Bias P and Bias N are represented in figure 8. The operational amplifier A1 3 comprises eleven bipolar transistors T7-T17. The non-inverting input of amplifier 3 corresponds to the base of an NPN transistor T7. Its inverting input corresponds to the base of an NPN transistor T8. Its output is taken on the emitter of a PNP transistor T9. The collector of transistor T7 is connected to terminal Vcc while its emitter is connected to the collector of an NPN transistor T10 whose emitter is grounded (terminal B). The emitter of transistor T7 is also connected to the base of a PNP transistor T11. The emitter of transistor T11 is connected, with the emitter of a PNP transistor T12, to the collector of a PNP transistor T13 whose emitter is connected to terminal Vcc. The base of transistor T12 is connected to the emitter of transistor T8 whose collector is connected to terminal Vcc and whose base constitutes the inverting input of amplifier 3. The emitter of transistor T8 is also connected to the collector of an NPN transistor T14 whose emitter is grounded. The collectors of transistors T11, T12 are respectively connected to the collectors of NPN transistors T15, T16. The bases of transistors T15, T16 are connected to the collector of transistor T15 and their emitters are grounded. The collector of transistor T16 is connected to the base of transistor T9 whose collector is grounded. The emitter of transistor T9, which constitutes the output of amplifier 3, is connected to the collector of a PNP transistor T17 whose emitter is connected to terminal Vcc. The bases of transistors T13 and T17 are connected to terminal Bias P. The operational amplifier A2 6 comprises five bipolar transistors T18-T22. The non-inverting input of amplifier 6 corresponds to the base of a PNP transistor T18. Its inverting input corresponds to the base of a PNP transistor T19 while its output corresponds to the collector of transistor T19. The emitter of transistor T18 is connected, with the emitter of transistor T19, to the collector of a PNP transistor T20 whose emitter is connected to terminal A. The collectors of transistors T18, T19 are respectively connected to the collectors of NPN transistors T21, T22. The bases of transistors T21, T22 are connected to the collector of transistor T21 and their emitters are grounded. The transconductance amplifier gm 9 comprises six bipolar transistors T23-T28. The non-inverting input of amplifier 9 corresponds to the base of a PNP transistor T23. Its inverting input corresponds to the base of a PNP transistor T24 while its output is taken on the collector of a PNP transistor T25 through a resistor R8. The emitter of transistor T23 is connected, with the emitter of transistor T24, to the collector of a PNP transistor T26 whose emitter is connected to terminal A. The collectors of transistors T23, T24 are respectively connected to the collectors of NPN transistors T27, T28. The bases of transistors T27, T28 are connected to their own collector while their emitters are grounded. The emitter of transistor T25 is grounded and its base is connected to the collector of transistor T24. A resistor R9 is connected between a resistor R8 and terminal A. The reference current generator 7 and the polarization generators 5 and 8 respectively correspond to PNP transistors T29, T30, T31. The emitters of transistors T29 and T31 are connected to terminal A while the emitter of transistor T30 is connected to terminal Vcc. The output of each generator 5, 7, 8 respectively corresponds to the collector of a respective transistor T30, T29, T31. The positive temperature coefficient current source 10 comprises an NPN transistor T32 whose emitter is grounded and whose collector is connected to the positive input M+ of the microphone amplified signal. The diodes D1 and D2 correspond to respective NPN transistors T33, T34 whose base and collector are interconnected. The circuit also comprises three transistors T35, T36, T37 for biasing the various transistors of the amplifiers 6 and 9. The emitter of a PNP transistor T35 is connected to terminal A while its collector is connected to the base of a PNP transistor T36 and to the collector of an NPN transistor T37. The emitter of transistor T37 is grounded. The emitter of transistor T36 is connected with the base of transistor T35 while the base of transistor T37 is connected to terminal Bias N. The bases of transistors T20, T26, T29, T30, T31 are connected together to the emitter of transistor T36 while the bases of transistors T10, T14 and T32 are connected to the base of transistor T37 (terminal Bias N). A resistor R10 is arranged between the inputs M+ and M- of the speech signal and a resistor R11 is arranged between base and emitter of transistor T3. The other elements appearing on figure 8 are designated by the same reference as in figure 1. Of course, many variants and modifications of the invention will appear to those skilled in the art. In particular, each of the disclosed components (operational amplifiers, transconductance amplifier, current sources, diodes, ...) can be replaced by one or a plurality of elements having the same function provided it remains in the scope of the claims. The current sources made by means of transistors could for example be made by means of resistors.

Claims: A transmit amplifier for a telephone set without self supply, comprising a LED (1) having an anode connected to a first positive terminal (A) of the telephone line and means for determining the ratio between the line current (IL) and the line voltage (VL), characterized in that it comprises supply means for supplying LED (1) with a constant current and deriving means comprising a transconductance amplifier (9) whose feed-back loop controls a switch (T3) that derives the current flowing in LED (1) when the voltage (VL) across the terminals (A, B) of the line becomes lower than a predetermined value (Vs1). The transmit amplifier of claim 1, wherein said supply means comprise a first operational amplifier (6) measuring the current (lied) in the LED (1) and whose feed-back loop limits this current (Iled) to a value (Is2) determined by a reference current generator (7). The transmit amplifier of claim 2, wherein the cathode of LED (1) is connected to the non-inverting input of said first amplifier (6). The transmit amplifier of claim 2 or 3, wherein said reference current generator (7) comprises a bipolar PNP transistor (T29) having an emitter connected to the first terminal (A) of the line and a collector connected to the inverting input of said first amplifier (6). The transmit amplifier of any of claims 1 to 4, wherein said switch comprises an NPN transistor (T3) whose collector is connected to the first terminal (A) of the line and whose emitter is connected, through a low value resistor (R3), to the cathode of the LED (1). The transmit amplifier of any of claims 1 to 5, wherein said deriving means also disconnect said supply means. The transmit amplifier of any of claims 1 to 6, wherein said means for determining the ratio between the line current (IL) and the line voltage (VL) comprise a second operational amplifier (3) whose inverting input receives the speech signal to be transmitted, added to a DC component of the line voltage (VL0) provided by a telephone exchange. The transmit amplifier of claim 5, wherein the feed-back loop of said second amplifier (3) comprises two Darlington-connected NPN transistors (T1, T2) and two diodes (D2, D1), the collector of the second transistor (T2) being connected to the inverting input of the transconductance amplifier (9) and the non-inverting input of the transconductance amplifier (9) being connected to the anode of the second diode (D2).

Inventors: DASGUPTA UDAY, ONG CHIN PUEH, DASGUPTA, UDAY, ONG, CHIN PUEH, DASGUPTA, UDAY, C/O SGS-THOMSON MICROELEC. PTE LTD, ONG, CHIN PUEH, C/O SGS-THOMSON MICROELEC. PTE LTD
Applicants: STMICROELECTRONICS PTE LTD, STMICROELECTRONICS PTE LTD.
Application year: 1994
Main classification: H04M1
IPC codes: H04M1