Slew rate control and optimization of power consumption in a power stage

Abstract: Power consumption by the driving circuitry of an output stage, employing a slew-rate controlling operational amplifier, is reduced by modulating the level of the current output by the operational amplifier in function of the working conditions of the output stage. Switching delay may also be effectively reduced. An auxiliary current generator (P 3 , P 4 ) forces an additional current through the conducting one of the pair of input transistors (N 1 , N 2 ) of the operational amplifier only during initial and final phases of a transition, essentially when the slew rate controll loop ceases to be effective. The boosting of the bias current through the conducting input transitor is determined by the degree of unbalance of the differential input stage of the operational amplifier, without the use of dissipative sensing element.

Description: The present invention relates to a method and relative driving circuit of an output power stage which permits an effective slew rate control and wherein current absorption by the driving circuitry is optimized in function of the operating conditions of the output power stage. The so-called intelligent power switches are finding widespread industrial application. These devices permit to drive any kind of loads (capacitive, inductive and/or resistive), referred to ground (in a high-side driver configuration) or to the power supply voltage (in a low-side driver configuration). They are called "intelligent" because on a single chip are integrated circuits capable of protecting and regulating in a real time mode the state of the integrated power switch (for example to protect the integrated power device from a short circuit etc...). At present, the most used technology for fabricating these integrated circuits is the so-called mixed of BCD technology (Bipolar-CMOS-DMOS). The integrated switch is often realized by a DMOS transistor, which, as compared with a bipolar transistor of similar current handling capacities, has the advantage of being voltage-driven, is capable of withstanding the maximum process voltage and has a lower saturation resistance. On the other hand, in the field of industrial applications, the problem of electromagnetic disturbances that may be produced by continuous fast switchings of relays and equipments is very important. In order to minimize the electromagnetic noise that is produced, it is important to realize switches that are capable of ensuring suitably slow slew rates, in practice switching fronts having a controlled slope. A common approach for switching-on and off a power transistor (power switch) with controlled switching fronts is that of loading and discharging the driving node of the transistor (the gate in the case of a field effect power transistor) with a constant current. Such a driving scheme, in case of a high-side driver employing a field effect power transistor (FET) is depicted in Figure 1. EP-A-0 398 170 discloses a driver circuit for a power DMOST employing a differential amplifier that operates at a relatively low quiescent current when the DMOST is on. When the DMOST is to be switched from off to on, the differential amplifier tail current is briefly raised to a substantially higher value in order to quicken the charging of the parasitic capacitance of the gate of the driven DMOST. The tail current differential is determined by geometrically define IC quantities. GB-A-2,257,854 discloses a drive circuit for an inductive load, wherein switch elements provide for a fast discharge path of the capacitor of the feedback differentiating circuit sensing the current flowing through the output load driver transistor EP-A-0 326 968 discloses a driving circuit of a power transistor driving a load, were the slew rate of the output voltage of the driving circuit is controlled by modulating the output current of an op-amp of the circuit in function of the load as sensed by a feedback loop. As well known to a skilled technician, such a solution does not permit to obtain perfectly controlled slew rates, because the slope of the switching front depends on the capacitance of the gate node which is not constant but assumes different values as the transistor passes from an off condition to a saturation condition and from the latter to a so-called linear working condition (i.e. an operating zone of its characteristic where the FET exhibits a resistive behaviour). Moreover, such a solution normally requires a circuitry capable of speeding up the passage from an off condition to a saturation condition at the beginning of a rising front and viceversa at the end of the descending front, and also for speeding up the passing from a saturation condition to a linear operating condition at the end of a rising front and viceversa at the beginning of a descending front, in order to maintain the turn-on and turn-off delays acceptably small. An alternative approach used for controlling the slew rate is based upon the use of a high gain operational amplifier, configured as an integrating stage, employing a capacitor C in its feedback line, as shown in Figure 2. Compared to the first solution, this alternative solution permits to obtain perfectly controlled switching fronts, because they are exclusively defined by the ratios between I ON /C and I OFF /C. On the other hand, the operational amplifier must be provided with an output stage capable of driving the gate of the power FET (for example a DMOS transistor) which may have a parasitic capacitance of a value that may be as low as ten odd pF and as large as several thousands pF and a broad band in order to prevent oscillations. Notwithstanding such a last approach may theoretically achieve a perfect slew rate control by suitably dimensioning the current generators I OFF and I ON and the bootstrap capacitance C (which may also be connected externally of the integrated circuit and therefore adapted according to need), it has the drawback of being unable to optimize also power consumption and the switching delays. For these reasons, this solution is scarcely employed in systems and equipments where power consumption of the control circuitry represents a non negligeable aspect and destined to function at a relatively high switching frequency. Moreover, in case of a power MOSFET, the driving must commonly be effected with a boosted voltage for reducing the resistance Ron of the power transistor. Therefore it is important that the current absorption from the supply node of the final stage of the driving operational amplifier, which is adequately boosted by employing a charge pump circuit, be as slow as possible to avoid overloading the charge pump circuit. Therefore there is the need and/or utility for an improved driving system which, beside allowing an easy slew rate control similar to that provided by a driving an integrating stage), also allows to optimize power consumption and to limit turn-on and turn-off delays. This objective is reached by the driving system of the invention which essentially consists in modulating the level of the current delivered by a driving operational amplifier and therefore current absorption, in function of different operating conditions of the output power transistor. At the base of the invention, there is the recognition of the fact that the current required for suitably drive the output power transistor depends on the zone of its characteristic in which the power transistor is functioning. In case of a MOSFET, when the transistor is completely on or off, the only current that is necessary to provide to its gate is just that sufficient to counterbalance the leakage and to ensure the actual state of the transistor (on or off). During rising and descending switching phases, the current required is equal to about: I = C GATE · V S T where C GATE is the capacitance seen from the gate node of the MOSFET, T represents the rise time or the fall time and V S is the supply voltage of the circuit. This current is such as to permit to the voltage swing of the output node to maintain a preset slope given by I ON,OFF /C. Moreover, in order to speed up the passing from an off condition to a saturation condition, at the beginning of a rising front, and viceversa, at the end of a descending front, and also for speeding up the passage from a saturation condition to a "linear" condition, at the end of a rising front, and viceversa, at the beginning of a descending front, the current delivered by the driving amplifier should be suitably increased during these phases of the operating cycle of the power switch in order to reduce switching delays. According to a first aspect of the invention which is defined by the claims, the current output by the driving operational amplifier and which is absorbed from the supply node of the final stage of the driving operational amplifier, is modulated as follows: a) the current is minimum when the operational amplifier is completely unbalanced, a condition that occurs when the output power transistor is in a ON or in a OFF state; b) the current is sufficient to ensure that the voltage on the output node of the circuit vary according to a preset slope for the entire duration of a rising or of a descending transition, during which the driving operational amplifier works in linear condition (under the control exerted by its own feedback loop) and the output power transistor is in saturation; c) the current is maximum when the output power transistor crosses in one sense or in the other, the boundary between the OFF region and its "linear behaviour" region (resistive behaviour) of its characteristic and the value of which has an influence on the turn-on and on the turn-off delay times. In practice, the system of the invention implements a modulation of the level of the current that is output by the driving operational amplifier in function of the operating conditions of a differential input stage of the operational amplifier itself, which reflect corresponding operating conditions of the output power transistor driven by the operational amplifier, as will be clearly explained and illustrated further on in this description. The modulation of the driving current delivered by the operational amplifier to the control node of the power transistor is implemented by a special circuit capable of selecting among different bias conditions of the differential input stage (that is of current absorption), in function of the state of unbalance of the same differential input stage. The differential input stage may be a transconductance stage having a differential current output, and a plurality of current mirrors implement a final, single-ended stage which drives the output power transistor. The different aspects and advantages of the invention will become more evident through the following description of several important embodiments and by referring to the attached drawings, wherein: Figure 1 is a basic circuit diagram for the driving of an output power transistor (high-side driver) as described above; Figure 2 is a basic circuit diagram for driving an output power transistor (high-side driver) by means of an integrating stage realized with a conventional operational amplifier, as described above; Figure 3 shows a driving circuit for an output power transistor (high-side driver), made according to the present invention; Figure 4 shows diagrams of the driving current delivered by the operational amplifier of Fig. 3; Figure 5 shows diagrams of significant voltages of the circuit of Fig. 3, for an entire cycle of turning-on and turning-off of the output power transistor; Figure 6 shows an input circuit for a logic switching signal, suitable to reduce the turn-on and switching signal, suitable to reduce the turn-on and turn-off delay times; Figure 7 is a diagram of the transfer function of the driving operational amplifier of Fig. 3, for a resistive load; Figure 8 shows a driving circuit made according to the present invention for an output power transistor configured as a low-side driver. A driving circuit according to the present invention for a power transistor PW 1 , configured as a high-side driver, is depicted in Fig. 3. Basically, a circuit of this type is capable of driving any type of load and the output node OUT of the circuit may also assume negative potentials (below ground). The latter characteristic is often required in driving heavy inductive loads (of several Henry), in order to speed up the demagnetization of the load. Without this specific requirement, the current mirror, mirror3 , could be referred to the ground potential instead of the output node potential, as in the shown example. The driving operational amplifier comprises three current mirrors ( mirror 1, mirror 2 and mirror 3) which substantially form a final stage, and a differential input stage that is essentially composed of the differential pair of transistors N 1 and N 2 and the respective current generators I Q . The differential input stage of the operational amplifier is a transconductance stage with a differential current output (implemented through the drain currents of the transistors N1 and N2). The three current mirrors: mirror 1, mirror 2 and mirror 3, form a final stage of the operational amplifier, the single-ended output of which may be directly connected to the gate of the output power transistor PW 1 , as shown. Of course, the mirror ratios of the second and third current mirrors: mirror 2 and mirror 3, which in the example shown are indicated as 1:10, may be of any suitable value, also a 1:1 ratio. Of course, a relatively large mirror ratio, as in the example shown, reduces current absorption in the differential input stage for the same level of output current. According to the method of the present invention, the output current delivered by the driving operational amplifier is modulated in function of the state of operation of the circuit, by the circuit arrangement implemented with the transistors M 1 and M 2 , the transistors P 1 , P 2 , P 3 and P 4 and with the auxiliary current generator I M . In general, all the current generators used in the circuit are not necessarily ideal generators but they are preferably realized with current mirror structures capable of delivering current as long as the voltage of their charge node remains close to the voltage to which they are referred (powered). The transistors M 1 and M 2 are cross-coupled to the differential input pair of transistors of N 1 and N 2 , and as the latter they are controlled by the signal present on the inverting input node in- of the operational amplifier and by the reference voltage V R applied to the noninverting input in+ of the amplifier, respectively. M 1 and M 2 have the function of limiting the output current of the driving operational amplifier to a level given by 2I Q , when the transconductance differential input stage is completely out of balance, that is when the signal present on the inverting input in- reaches ground potential or the potential 2V R . In fact, when the voltage on the input node indrops below zero, the output transistor PW 1 is on and its gate is biased at the V CP voltage, which may be a boosted voltage generated by a charge pump circuit (not shown in the figures). Commonly, in order to suitably overdrive a power MOSFET, the boosted voltage V CP is about 10V higher than the supply voltage V S , in order to optimize the internal resistance R DSon of the power transistor. In these conditions of operation, the operational amplifier, that is its differential input stage, is completely out of balance. In a state of full unbalance, in the above reiterated sense, of the differential input stage, the transistors M 1 and P 2 are on, while M 2 , P 1 , P 3 and P 4 are off. Therefore the transistor N 2 , through its collector, absorbs from the boosted voltage node V CP a current given by 2I Q . This condition persists for the full range of variation of the voltage on the inverting input node in- comprised between zero and the voltage V TH1 , given by: V TH 1 = V R - V BE ( N 2 ) - V TH ( M 1 ) where V TH1 is the threshold voltage of the transistor M 1 . A dual situation occurs when the output power transistor PW 1 is off. In this case, the inverting input in- of the driving operational amplifier is at the voltage 2V R . As a consequence M 2 and P 1 are on while M 1 , P 2 , P 3 and P 4 are off and in this case the transistor N 1 absorbs through its collector a current given by 2I Q . This condition persists for the whole range of values of the voltage on the inverting input node in- that are comprised between 2V R and the voltage V TH2 , given by: V TH 2 = V R + V BE ( N 1 ) + V TH ( M 2 ) where V TH2 is the threshold voltage of the M 2 transistor. Clearly the current generators I Q may be sized so as to force a minimum current, that is a current that is just sufficient to compensate for dispersion currents of the gate node of the output power transistor PW 1 and therefore maintain a conducting state or a cut-off state. During a rising or descending transient (turn-on or turn-off of the power transistor), when the driving operational amplifier works in a linear manner, that is when the voltage on the inverting input node in- of the driving operational amplifier remains the vicinity of the reference voltage V R , as diagrammatically depicted in Fig. 4, the feedback loop implemented by the capacitance C, assumes the control of the rise or fall slew rate thus ensuring a pre-established slope and the operational amplifier delivers the current that is necessary to charge or discharge the gate node of the output power transistor PW 1 . In fact, in these conditions of operation, the collector currents I C (N 1 ) and I C (N 2 ) are tied to the voltages present on the inverting and on the noninverting inputs of the operational amplifier by the following trascendental relationships (corresponding to the exponential zones of the current diagrams of Fig. 4): At the beginning and at the end of a rise transient, respectively, the output power transistor passes from an off condition to a condition of linear operation and from the latter to a saturation condition. Similarly, at the beginning and at the end of a fall transient, the output power transistor passes from a saturation condition to a linear operation condition and from the latter to an off condition, respectively. During these phases, a relatively large charge or a discharge current of the driving node of the power transistor is required, for example in order to quickly reach the turn-on threshold V TH of the power transistor on turning on and at the end of the turn-on transient, in order to quickly bring the V GS voltage of the output power transistor to about 10V in order to optimize the resistance R DSon . According to the present invention, such a momentary requirement for a relatively high output current of the driving operational amplifier is satisfied by allowing the respective transistor of the differential pair of the input stage of the operational amplifier to absorb a markedly larger current than the curren 2I Q that is fixed by the two current generators I Q , through a dedicated auxiliary current generator I M . This auxiliary current generator I M is designed so as to force through the collector of the transistor that is conducting of the input differential pair of transistors, an additional current I M , which is much greater than the current 2I Q forced by the two current generators I Q that bias the differential input stage. This is obtained by momentarily turning on one or the other of the two transistors P 3 and P 4 , during a phase of relatively small unbalance of the differential input stage. The explanation that follows will be more easily understood by referring also to the voltage diagram of Fig. 5. In a turn-on phase of the output power transistor PW 1 , the state of relatively small unbalance of the differential input stage of the operational amplifier coincides with the period during which the feedback loop of the operational amplifier is inactive, because the voltage on the gate node of the output power transistor has not yet risen above the turn-on threshold V TH . As a consequence it is the transistor P 3 that turns on. Once the output voltage of the driving operational amplifier becomes greater than the threshold voltage V TH , the output power transistor starts to conduct and therefore the feedback loop that controls the operation of the driving operational amplifier (the slew rate of the output voltage V out ) in the vicinity of the reference voltage V R is activated for the duration of the rise transient. At the end of the rise transient, that is when the output voltage V out of the driving operational amplifier reaches the limit value, given by V S +V GSon , the voltage on the feedback capacitor C ceases to rise and also in this case the function of the feedback loop that controls the slope of the switching front (slew rate) ceases. At this point, the transistor P 3 turns on again, thus allowing again the current forced by the driving operational amplifier (that is by the differential input stage) to rise to the maximum value, given by I M +2I Q . In this way, the reaching of a full turn-on condition (overdrive) of the output power transistor is accelerated by quickly charging its control gate to the boosted voltage V CP , which may be generated by a suitable charge pump circuit according to a common practice. The turn-off of the output power transistor PW 1 occurs in a reverse manner. Also in a turn-off phase, by the turning on of the transistor P 4 , the maximum current given by the sum of: I M +2I Q , is forced by the driving operational amplifier during a discharge of the gate node of the output power transistor from the overdrive voltage V CP to the voltage given by V S +V GSon . At this point, the feedback loop that controls the slope of the fall front (slew rate) becomes active as long as the turn-off threshold V TH of the output power transistor is overcome, which disactivates the feedback loop and causes again the turning-on of the transistor P 4 and thus the forcing by the driving operational amplifier of the maximum current until the voltage 2V R is reached by the inverting input node V(in-). By referring to Fig. 5, during a turn-on phase of the output power transistor PW 1 , the delay T 1 corresponds to time necessary for the inverting input node (v(in-)) to discharge, because of the turning-on of the current generator I ON and the turning-off of the current generator I OFF , which are controlled by a pair of phase-opposed, logic signals (input and input\), during the drop of the v(in-) voltage from 2V R down to V R . The delay T 2 represents the time necessary for the gate potential of the output power transistor to charge from ground potential up to a level beyond the turn-on threshold voltage V TH . It is this second contribution T 2 to the turn-on delay (and similarly also to the turn-off delay) that is advantageously reduced in the circuit of the invention by allowing the driving operational amplifier to deliver a relatively high current for quickly charging the gate node, during these phases. Therefore the turn-on delay that is given by the sum of the delays T 1 and T 2 is markedly reduced. A similar reduction of the total delay time is achieved during a turn-off phase. On the other hand, by analizing the first of the two delay factors, it may be observed that the time T 1 necessary for the inverting input of the operational amplifier to reach the reference voltage V R (starting from 2V R on turning-on or from 0 on turning-off) is given by: T 1 = V R · C I ON,OFF where C is the feedback capacitance that controls the slew rate. Also such a first contribution to the switching delay (turn-off delay and turn-on delay) may be effectively reduced by employing a special input circuit as the one depicted in Fig. 6. By employing an input circuit as the one depicted in Fig. 6, the so optimized delay T 1opt , is given by the following expression: T opt = V BE · C I ON,OFF In fact, with the input circuit of Fig. 6, the inverting input (in-) of the driving operational amplifier assumes istantaneously a V BE voltage. Therefore, beside optimizing the T 2 contribution to the turn-on delay (and to the turn-off delay), also the T 1 may be markedly reduced by employing an input circuit as the one depicted in Fig. 6. Of course, it is important to ensure stability of the circuit. In the example shown, the mid-band gain of the feedback loop is given by: G 0 = G m · R G · g m · R L 1 + g m · R L where R G is the resistance seen from the gate node, g m is the transconductance of the output power transistor PW 1 , R L is the load resistance and G m is given by the following relationship: G m = 20 2 · V T I E + R 1 + V T I E - I Q where I E may be derived from the following trascendental relationship: R 1 · ( I E - I Q ) = V T · ln I Q I E - I Q Resistances R 1 and R 2 of an appropriate value may be introduced (if necessary) in order to ensure stability of the circuit. Their effect is to make less steep the exponential rise and fall curves of the collector currents I C (N1) and I C (N2) in their respective diagrams shown in Fig. 4. Basically the feedback capacitor C introduces a "zero" at the origin and a "pole" of value p 1 =1/R V C, where R V is the resistance seen by the capaciter C. The resistance R V is given by the sum of two contributions. The load resistance of the circuit R L (not shown in the circuit diagrams) and the resistance seen from the inverting node in- of the driving operational amplifier. Such a pole is the lowest because it is the one that determines the slew rate. A second pole is introduced by the parasitic capacitance of the gate node of the output power MOSFET PW 1 and has a value given by p 2 =1/C G R G . Other poles at higher frequencies are introduced by the operational amplifier. Therefore in designing the driving operational amplifier it is required that the pole p 2 remains dominant as compared with all the other poles. For a purely resistive load, the transfer function is graphically depicted in Fig. 7. Of course, if the load is not purely resistive, the poles and zeroes that are introduced concur to modify the transfer function and therefore it is necessary to verify that the stability requirements be satisfied, case by case. Of course the invention may be applied also in the case of a configuration of the output power transistor as a low-side driver. Such an alternative embodiment of the invention is depicted in the circuit diagram of Fig. 8, for which the same considerations that have been made above for the case of a high-side driver configuration, hold. According to preliminary CAD simulations, the driving circuit of the invention permits to readily achieve turn-on and turn-off delay of about 400 nanoseconds with a slew-rate of about 5V/µsec.

Claims: A method of switchingly driving an output stage constituted by a power transistor (PW 1 ) by way of a driving operational amplifier (I Q , N 1 , N 2 , mirror1, mirror2, mirror3), characterized in that the slew rate of the voltage (OUT) on the output node of the output stage (PW 1 ) during rising and descending transients is controlled by modulating the level of the current (I c (N 1 ), I c (N 2 )) output by said driving operational amplifier in function of the passing from a condition of saturation to a condition of linear functioning of said transistor (PW 1 ) of the output stage as sensed by way of a feedback loop. A driving method as defined in claim 1, wherein said transistor of the output stage is a field effect power (PW 1 ) transistor and the method comprises delivering a driving current of a preset minimum value (I OFF ) when said transistor (PW 1 ) is on or off and said driving operational amplifier is fully out of balance; delivering a driving current of a preset maximum value (I ON ) when said transistor passes, in either direction, from an off region to a linear operation region of its working characteristic and said operational amplifier is in a state of limited unbalance; delivering a driving current controlled by a feedback loop (C) of the operational amplifier so that the voltage of the output node (OUT) of the output stage varies with a preset slew rate during rising and descending t.ransients and while said field effect power transistor (PW 1 ) is in saturation. A driving method according to claim 1, wherein the modulation of the level of the current (I C (N 1 ), I C (N 2 )) output by said driving operational amplifier is implemented by controlling the collector current of a transconductance, differential input stage (N 1 , N 2 , I Q ) of said driving operational amplifier I Q , N 1 , N 2 , mirrorl, mirror2, mirror3). A driving method according to claim 3, wherein during conditions of operation of said output stage that disable the control of the output current exerted by the feedback loop of the driving operational amplifier, the modulation of the level of the output current is realized by enabling and disabling an auxiliary current generator (I M ) forcing a certain current through the conducting transistor of said pair of transistors, depending on the state of unbalance of said differential input stage. A driving circuit for an output stage, employing a driving operational amplifier comprising a transconductance, differential input stage (N 1 , N 2 , I Q ) and a current mirror final stage (mirrorl, mirror2, mirror3), characterized by comprising a feedback capacitor (C) connected between an output node (OUT) and an inverting input node (in-) of said differential input stage, an auxiliary current generator (I M ) capable of forcing through the conducting one or a pair of input transistors (N 1 , N 2 ) of the differential input stage a current markedly larger than the current forced by respective biasing current generators (I Q ) of a pair of input transistors (N 1 , N 2 ) of said differential input stage; switching means (P 3 , P 4 ) controlled by a second pair of transistors (M 1 , M 2 ), cross-coupled with the inputs (in+, in-) of said differential input stage, capable of enabling and disabling current paths between each of said transistors (N 1 , N 2 ) of said differential input pair and said auxiliary current generator (I M ). A driving circuit as defined in claim 5, wherein the transistors (N 1 , N 2 ) of said differential input pair are bipolar NPN transistors, each having a collector cornected to an input of a current mirror (mirror1, mirror2). A driving circuit as defined in claim 6, wherein the transistors of said second pair (M 1 , M 2 ) are field effect transistors, each having a gate connected to a respective input (in-, in+) of said differential input stage, a source connected to the emitter of the NPN transistor (N 2 , N 1 ) of the other input and a drain connected in common to a respective current generator (I Q ) and to the base of a pair of PNP transistors (P 3 , P 4 ), the first of which has a collector connected to said auxiliary current generator (I M ) and an emitter connected to the emitter of the NPN transistor (N 2 ) of said input (in+) while the other PNP transistor has a collector connected to said auxiliary current generator (I M ) and an emitter connected to the emitter of the other NPN transistor (N1)of said other input (in-). A driving circuit according to claim 7, wherein between the emitter of the PNP transistors (P 3 , P 4 ) of said pair and the emitter of the respective NPP input transistor there is a stabilizing resistance (R 1 , R 2 ). A driving circuit as defined in claim 5, characterized by comprising an input circuit (2·V R , I OFF , I ON ) for a switching logic signal ( input ), composed of a complementary stage which comprises a first NPN transistor, a second n-channel field effect transistor, a third p-channel field effect transistor and a fourth bipolar PNP transistor, functionally connected in series with each other between a node maintained at a voltage (2V R ) that is twice the value of the reference voltage (V R ) that is supplied to the non inverting input (in+) of said driving operational amplifier and a common ground node, the base of said first NPN transistor and of said fourth PNP transistor being connected in common to said reference voltage (V R ), the switching logic signal ( input ), being applied to the gates connected in common of said second and third field effect transistors, the emitters of which are connected in common to the non inverting input node (in-) of the driving operational amplifier.

Inventors: BONTEMPO GREGORIO, GARIBOLDI ROBERTO, PULVIRENTI FRANCESCO, BONTEMPO, GREGORIO, GARIBOLDI, ROBERTO, PULVIRENTI, FRANCESCO
Applicants: CONS RIC MICROELETTRONICA, ST MICROELECTRONICS SRL, CONSORZIO PER LA RICERCA SULLA<BR>MICROELETTRONICA NEL MEZZOGIORNO - CORIMME, STMICROELECTRONICS S.R.L.
Application year: 1994
Main classification: H03G1
IPC codes: H03G1, H03K17