Low distortion circuit with switched capacitors

Abstract: A switched capacitor circuit (1) comprising an operational amplifier (OA), having first and second input terminals and an output terminal, the first input terminal being connected to a first reference potential, said operational amplifier (OA) being provided with a negative feedback network including a first capacitive element (C1) which is connected between the second input terminal and the output terminal of the operational amplifier, a second capacitive element (C2) which has a first terminal alternately connected to the second input terminal of the operational amplifier (OA) and to a reference potential, and a second terminal connected to a first circuit node (A) which is alternately connected to a signal input terminal (VIN) and said first output terminal (VOUT) of the operational amplifier (OA), the circuit (1) further comprising a third capacitive element (CX) connected between said circuit node (A) and a reference potential.

Description: This invention relates to switched capacitor circuits, in particular to a switched capacitor filter with reduced harmonic distortion. A switched capacitor circuit, also referred to briefly as SC circuit in literature, is a circuit capable of processing sampled analog signals using clock signals and switching circuit means driven by such signals. An example of an SC circuit is the SC integrator shown in Figure 1, which constitutes a basic circuit block for obtaining an SC filter. In this case, the four switching means of the circuit, denoted by SW1, SW2, SW3 and SW4 in the drawing figure, are electronic switches implemented with MOS transistors, and are driven each by two discrete non-overlapping phase clock signals. In principle, there should occur no alterations to the charge stored in the capacitors of the SC circuits, and accordingly, the charge transfers should be exact ones. However, due to the presence of parasitic capacitances associated with the channels of the MOS transistors used as switching means, this is not what precisely happens in actual practice. In fact, these parasitic capacitances store an amount of charge which is proportional to the signal that appears at the terminals of the respective switching means, and when these charges are injected into the capacitors, the event produces a voltage offset in the system if said signal is a constant (DC) one, or an error in gain and an amount of harmonic distortion in the output signal if said signal is a varying one. Several methods are described in literature for reducing this cause of harmonic distortion in switched capacitor circuits, although it is stated by D. G. Haigh and B. Singh in an ISCAS 1983 paper "A switching scheme for switched capacitor filters which reduces the effect of parasitic capacitances associated with switch control terminals" that in SC circuits with two non-overlapping phase clock signals, said harmonic distortion can never be brought below a definite level, calculated at around 40-50 dB. This is indeed due to the presence of said parasitic capacitances associated with the channel of the MOS transistors employed as the switching means. The solution proposed in the report mentioned above for reducing said harmonic distortion to a minimum, consists of using four discrete clock signals with non-overlapping phases, a method known to the skilled ones in the art as "four-phase clocking". EP-A-0 439 276 discloses a switched-capacitor circuit having means for reducing signal-dependent distortion. In another paper ("On switch-induced distortion in switched capacitor circuits", ISCAS 1988), D. G. Haigh and J. T. Taylor state that in switched capacitor filters, harmonic distortion never drops below values in the -68 dB range, not even when a four clocking signals system is used. To solve the problem, it is proposed therein of increasing the transition time for the phases of the clocking signals to change from one state to the other. However, if SC filters are to be designed with even lower harmonic distortion levels, the last-mentioned approach is no longer proposable because it would involve a very long transition time. In the circuit shown in Figure 1, which is an SC integrator of conventional construction, the electronic switches shown at SW1, SW2, SW3 and SW4 in the drawing figure may each comprise, for example, a pair of n-channel MOS transistors, or pair of p-channel transistors, or pair of CMOS transistors. The latter choice is the most comprehensive one, and an obligatory one for values of the input/output signals comparable with the power supply ones. The two transistors jointly making up a switch are driven by two clocking signals which do not overlap in time, such as the signals F1 and F2 and the signals F1a and F2a, whose waveforms are shown in Figure 2. For an analysis of the effect of the charges stored in the channel by the MOS transistors, one should consider the input circuit of the integrator, shown best in Figure 3, which comprises a capacitor C3 and the two switches SW3 and SW4. Similar considerations would apply to the other switched capacitor C2 being feedback connected, together with the switches SW1 and SW2, to the operational amplifier included in the integrator. The two n-channel MOS transistors, M1 and M2, which make up the switch SW3, are driven by the clocking signals F1 and F2, with the two transistors M3 and M4 of the switch SW4 being driven by the signals F1a and F2a. As shown in Figure 2, the signal F1a changes in level before the signal F1, and the signal F2a changes in level before the signal F2. The charges stored in the channels of the transistors M1 and M2, in the conduction state, of the switch SW3 are not injected into the capacitor C3 because, upon such transistors being cut off by the drive signals F1 and F2, the other end of the capacitor C3 is floating, in that the signals F1a and F2a are at a low already. The charges which are stored into the transistors M3 and M4 of the switch SW4 will now be considered. Upon M3 being cut off, the charge stored in its channel is partly injected into the capacitor C3. This amount of charge is. however, constant at all times, because the voltage across M3 is fixed (virtual ground of an operational amplifier) and the other end of the capacitor C3 is also connected to a fixed (ground) voltage reference via the transistor M2, which transistor is still in a conduction state. This constant charge injection produces a voltage offset in the output signal but no harmonic distortion. The charge stored up in the transistor M4, which, in the transition to the cutoff state from the conduction state is injected into the capacitor C3, is a non-constant quantity, despite the voltage across M4 being a fixed (ground) one, because the other end of the capacitor C3, connected to a circuit node denoted by B in Figure 3, is at the input voltage VIN. The circuit node B sees, toward ground, an intrinsic parasitic capacitance Cp which is due to the source and drain diffusions of the two transistors M1 and M2, and a conduction resistance RON of the transistor M1. As those familiar with the art will readily recognize, both the capacitance Cp and the resistance RON have a non-linear pattern versus voltage. Since the voltage at the circuit node B varies with the voltage of the input signal VIN, it follows that the impedance seen toward ground from the circuit node B, and consisting of a parallel of the parasitic capacitance Cp and the conduction resistance RON of the transistor M1, will also vary with the input voltage. This causes the amount of the charge injected into the capacitor C3 by the transistor M4 to be non-constant and vary with the input signal Vin, thereby adversely affecting the system linearity and introducing harmonic distortion in the output signal. It is the object of this invention to provide SC circuits having a lower harmonic distortion than prior art SC circuits, without significantly increasing the circuit complexity and the area required for integration as an integrated circuit. The advantages of an SC circuit according to the invention which is defined in claims 1 and 5 will be apparent from the description of an embodiment thereof, given by way of example and not of limitation with reference to the accompanying drawings. In the drawings: Figure 1 is a diagram of a prior art SC circuit; Figure 2 shows the waveforms of four non-overlapping phase clocking signals as commonly used with circuits of the kind shown in Figure 1; Figure 3 shows a detail of the circuit diagram in Figure 1, wherein the switching circuit means are n-channel field-effect transistors; Figure 4 shows a first embodiment of an SC circuit according to the invention; Figure 5 shows a second embodiment of an SC circuit according to the invention; Figure 6 shows a third embodiment of an SC circuit according to the invention; Figure 7 shows a detail of the circuit diagram in Figure 6, wherein the switching circuit means are field-effect transistors; and Figure 8 is a graph showing the values of the equivalent resistances of some circuit components plotted against voltage. The inventive circuit as shown in Figures 4, 5 and 6, does solve the above problem and offers significantly less harmonic distortion. The circuit diagram of Figure 7, which represents the input circuit for a circuit as shown in Figure 6, has, compared with the conventional circuit diagram of Figure 3, a capacitor CY connected between the circuit node B and ground. This capacitor CY is in parallel with the impedance seen toward ground from the circuit node B. By suitably sizing the capacitor CY so that its capacitance is much higher than the parasitic capacitance Cp of the node B and that its equivalent impedance is much lower than the conduction resistance Ron of the transistors at a frequency equal to the inverse of the decay time of the clocking signals used to drive the switch transistors, the overall impedance as seen from the circuit node B toward ground can be linearized. A typical value of the parasitic capacitance at the node B, due to the source and drain diffusions of the transistors M1 and M2, would be Cpo=0.1pF. This capacitance varies non-linearly with the voltage applied across it in accordance with the empiric formula, Cp = Cpo/((1+V/0.8)^0.38). The equivalent resistance associated with said capacitance is given by the formula, Reqp = 1/((2PI/tf)*Cp) where, tf is the decay time of the clocking signals used for driving the switch transistors, which time may be 3 ns, for example. The pattern of the resistance Reqp vs. voltage is illustrated by the graph of Figure 8. Also shown in that graph are the pattern of the conduction resistance Ron of the MOS transistors employed and that of the equivalent resistance of the added capacitance CY at the node B. It can be seen that, for a capacitance CY of 0.4 pF, the equivalent resistance is much smaller than the other two resistances, and what is more, linear with voltage. The equivalent resistance of the capacitor CY is given by the formula, Reql = 1/((2PI/tf)*CY) which, as can be seen, is unrelated to voltage. The overall equivalent resistance at the node B is given by a parallel of the three resistances plotted on the graph of Figure 8. The above resistance closely approximates, both in value and pattern vs. voltage, the smallest of the three resistances, namely the equivalent resistance of the capacitor CY. As previously explained in connection with the conventional input circuit of Figure 3, the equivalent resistance that the node B sees toward ground only becomes influential, as concerns circuit distortion, upon the transistor M4 passing from the conduction state to the cutoff state. The charges which have been built up in the transistor channel are, in fact, partly injected into the capacitor C3 in a proportional amount to the impedance present at the other end of the capacitor. When the other end of the capacitor, connected to the circuit node B, sees toward ground a resistance which is unrelated to the voltage at the node, the amount of injected charge is always a constant one unrelated to the input signal voltage. This constant charge injection causes a voltage offset in the output signal, but has no bearing on the system harmonic distortion. Similar considerations would also apply to the other two examples of an SC circuit embodying this invention, shown in Figures 4 and 5. Thus, the circuit of this invention does solve the aforementioned technical problem, and obtains a number of advantages, outstanding among which is that the effects from the parasitic capacitances, associated with the channel of the MOS transistors serving as electronic switches in circuits of this kind, can be significantly reduced and very low distortion, typically below -80 dB, SC circuits be provided. A further advantage is that the circuit complexity and the area required for integration of the circuit are kept practically unaltered. It should be understood that many modifications, adaptations, integrations, changes, and replacements of elements with functional equivalents thereof may be made unto the embodiments described hereinabove by way of examples and not of limitation without departing from the protection scope of the following claims.

Claims: A switched capacitor circuit (1) comprising at least one operational amplifier (OA), having at least first and second input terminals and at least a first output terminal, the first input terminal being connected to a first reference potential, said operational amplifier (OA) being provided with at least one negative feedback network including a first capacitive element (C2) which has a first terminal alternately connected, through a first switching circuit means (SW1), to the second input terminal of the operational amplifier (OA) and said first reference potential, and a second terminal connected to a first circuit node (A) which is alternately connected, through a second switching circuit means (SW2), to a signal input terminal (VIN) and said first output terminal (VOUT) of the operational amplifier (OA), the circuit (1) being characterized in that it comprises at least a second capacitive element (CX) connected between said first circuit node (A) and a second reference potential. A switched capacitor circuit as claimed in Claim 1, characterized in that said second reference potential is the same as said first reference potential. A switched capacitor circuit as claimed in either Claim 1 or 2 and being monolithically integrated to a substrate of a semiconductor material, characterized in that the capacitance of said second capacitive element (CX) is higher than the intrinsic capacitance of the integrated structure as seen from said first circuit node (A) toward the first reference potential. A switched capacitor circuit as claimed in any of the preceding claims, characterized in that the equivalent resistance of said second capacitive element (CX) is lower than the conduction resistance (Ron) of the second switching circuit means (SW2). A switched capacitor circuit (1) comprising at least one operational amplifier (OA), having at least first and second input terminals and at least one output terminal, the first input terminal being connected to a first reference potential, the circuit comprising a first capacitive element (C3), having a first terminal alternately connected, through a first switching circuit means (SW4), to the second input terminal of the operational amplifier (OA) and said first reference potential, and a second terminal connected to a first circuit node (B) which is alternately connected, through a second switching circuit means (SW3), to a signal input terminal (VIN) and to a second reference potential, the circuit (1) being characterized in that it comprises a second capacitive element (CY) connected between said first circuit node (B) and a third reference potential. A switched capacitor circuit as claimed in Claim 5 and being monolithically integrated to a substrate of a semiconductor material, characterized in that the capacitance of said second capacitive element (CY) is higher than the intrinsic capacitance of the integrated structure as seen from said first circuit node (B) toward the first reference potential. A switched capacitor circuit as claimed in any of Claims 5 and 6, characterized in that the equivalent resistance of said second capacitive element (CY) is lower than the conduction resistance (Ron) of the second switching circuit means (SW2). A switched capacitor circuit (1) as claimed in claim 5, characterized in that said operational amplifier (OA) is provided with at least one negative feedback network comprising a third capacitive element (C2) which has a first terminal alternately connected, through a third switching circuit means (SW1), to the second input terminal of the operational amplifier (OA) and said first reference potential, and a second terminal connected to a second circuit node (A) which is alternately connected, through a fourth switching circuit means (SW2), to a second reference potential and the output terminal (VOUT) of the operational amplifier (OA), and comprising a fourth capacitive element (CX) connected between said second circuit node (A) and a fourth reference potential. A switched capacitor circuit as claimed in any of Claims 5,6, 7 and 8, characterized in that said second reference potential is the same as said first reference potential. A switched capacitor circuit as claimed in any of Claims 5, 6, 7 and 8, characterized in that said third reference potential is at least the same as one of said first and second reference potentials. A switched capacitor circuit as claimed in Claim 8, characterized in that said fourth reference potential is at least the same as one of said first and second reference potentials. A switched capacitor circuit as claimed in any of Claims 8, 9, 10 and 11 and being monolithically integrated to a substrate of a semiconductor material, characterized in that the capacitances of said second and fourth capacitive elements (CY, CX) are higher than the intrinsic capacitances of the integrated structure as seen from said first and second circuit nodes (B and A) toward said third and fourth reference potentials, respectively. A switched capacitor circuit as claimed in any of Claims 8, 9, 10, 11 and 12, characterized in that the equivalent resistances of said second and fourth capacitive elements (CY, CX) are lower than the conduction resistances (Ron) of the second (SW3) and fourth (SW2) switching circuit means. A switched capacitor filter, characterized in that it comprises at least one switched capacitor circuit as claimed in any of the preceding claims.

Inventors: NAGARI ANGELO, NICOLLINI GERMANO, NAGARI, ANGELO, NICOLLINI, GERMANO
Applicants: ST MICROELECTRONICS SRL, STMICROELECTRONICS S.R.L.
Application year: 1994
Main classification: H03H19
IPC codes: H03H19, H03F3