Voltage drop circuit to be built in semiconductor IC chip

Abstract: A voltage drop circuit to be built in a high integrated IC chip, including a voltage divider (10a) and a voltage comparator (11). A divided output voltage V INT of the voltage divider and a comparison reference voltage V REF are applied to respective input terminals (-) and (+) of the voltage comparator. An output signal of the voltage comparator (SG11) is supplied to a control signal generator (10b) section of the voltage divider (10a) to generate control pulses (₁,₂) and to control the connection of capacitors of a voltage divider section of the voltage divider by the control pulses to supply the divided output voltage V INT to a load circuit (Z). In this case, the capacitors are used as a voltage dividing member for generating a voltage for an internal circuit from an external power source voltage. Thus, a power loss can be reduced and utilization efficiency of the capacitors is raised. Hence, the voltage drop circuit is suitable for high integrated IC chip.

Description: BACKGROUND OF THE INVENTION The present invention relates to a voltage drop circuit including a voltage divider circuit by using capacitors, and more particularly to a voltage drop circuit of a low consumption power type of this kind suitable to be built in a large capacity semiconductor dynamic random access memory (DRAM) chip. Description of the Related Arts A semiconductor IC chip such as a large capacity DRAM or a built-in voltage drop circuit for dropping a power source voltage from outside so as to operate internal circuits such as a memory cell array at a low voltage. A typical example of a voltage drop circuit of this kind in the prior art has a structure of a so-called series regulator configuration, as disclosed in a paper of the IEEE Journal of Solid-state Circuits, Vol. 25, No. 4, pp. 903-911, 1990. This conventional voltage drop circuit 120 is provided with a voltage comparator 121 and a p-type FET (field effect transistor) 122 to input an output of the voltage comparator 121 at its gate electrode, as shown in Fig. 1. A comparison reference voltage V REF is applied to one input terminal (-) of the voltage comparator 121 and a power source voltage V DD is supplied to the source electrode of the FET 122 via a power source terminal 124. And a voltage V INT output from the drain electrode of the FET 122 is fed to another input terminal (+) of the voltage comparator 121 and an output terminal 125. Between the output terminal 125 and an earth voltage point, a load circuit Z such as a memory circuit of the aforementioned semiconductor IC chip with a built-in voltage drop circuit 120 is connected. By applying the comparison reference voltage V REF and the output voltage V INT to the respective input terminals (-) and (+) of the voltage comparator 121, it is controlled so that a current flowing in the FET 122 may increase when the output voltage V INT drops rather than the comparison reference voltage V REF , so as to obtain a low voltage supply circuit to the load circuit Z. Another example of a conventional voltage drop circuit is disclosed in Japanese Patent Laid-Open No. Sho 63-121467, as shown in Fig. 2. In this case, as shown in Fig. 2, the voltage drop circuit 130 has the above-described series regulator configuration. That is, in addition to a combination of a comparator circuit 131 and an output FET 132 for inputting its output at the gate electrode, there is provided a capacitor voltage divider circuit 134 inserted between a power source terminal 135 of a power source voltage V DD1 and the source electrode of the output FET 132 and one input terminal (-) of the comparator circuit 131 is connected to the earth voltage point. The voltage divider circuit 134 is comprised of one series connection circuit which includes a first capacitor C131 coupled with the power source terminal 135 via a first switch S131, a second switch S132 and a second capacitor C132 connected to an earth terminal 136 of an earth voltage GND, another series connection circuit which includes a third switch S133 coupled with a connection point between the first switch S131 and the first capacitor C131 and a third capacitor C S connected to the earth terminal 136, a fourth switch S134 for selectively connecting the connection point between the first capacitor C131 and the second switch S132 with the earth terminal 136, and a fifth switch S135 for selectively coupling the connection point between the second switch S132 and the second capacitor C132 with the connection point between the third switch S133 and the third capacitor C S . A partial output voltage V DD3 obtained at the connection point between the third switch S133 and the third capacitor C S is applied to the source electrode of the output FET 132. An ouput voltage V DD2 obtained from the drain electrode of the output FET 132 is applied to the another input terminal (+) of the comparator circuit 131 and a low voltage operational load circuit Z via an output terminal 133. In this case, the first and second switches S131 and S132 and the third, fourth and fifth switches S133, S134 and S135 are opened or closed in response to first and second control signals Φ1 and Φ2, respectively. Next, the operation of the conventional voltage drop circuit 130 will now be described with reference to Figs. 3a and 3b which show an equivalent circuit of the voltage divider circuit 134 and its operational waveform view, respectively. When the first control signal Φ1 is a high level for the first and second switches S131 and S132 to be on and the second control signal Φ2 is a low level for the third, fourth and fifth switches S133, S134 and S135 to be off (the state shown in Fig. 2), the first and second capacitors C131 and C132 are connected in series between the power source voltage V DD1 and the earth voltage GND and become a charged state (a charge in Fig. 14a). At this time. to the output FET 132, the partial output voltage V DD3 corresponding to a holding capacity of the third capacitor C S is supplied and the output voltage V DD2 of the output FET 132 is fed to the load circuit Z via the output terminal 133. Next. when the first control signal Φ1 is the low level with the first and second switches S131 and S132 to be off and the second control signal Φ2 is the high level with the third, fourth and fifth switches S133, S134 and S135 to be on. the first and second capacitors C131 and C132 are cut from the power source voltage V DD1 and the first, second and third capacitors C131. C132 and C S are connected in parallel with each other (a discharge in Fig. 3a), and the total capacity of the first, second and third capacitors C131, C132 and C S . which is the maximum voltage V DD3 max is inserted between the source electrode of the output FET 132 and the earth voltage point 136. In the high level period of the second control signal Φ2. the output voltage V DD2 of the FET 132 drops from the whole capacity V DD3 max, that is, (1/2) · V DD1 to the level of (1/2) · V DD1 - ΔV with the passage of discharge time, and in the high level period of the first control signal Φ1, including the whole switch non-continuity period, since the output voltage V DD2 becomes only the holding capacity of the third capacitor C S . its level further drops from the (1/2) · V DD1 - ΔV (Fig. 3b). In the first conventional voltage drop circuit shown in Fig. 1, a power loss P (V DS × I DS ) due to a voltage V DS and a current I DS between the source and drain electrodes of the FET 122 is caused to bring about a rise in temperature of the IC chip constituting the load circuit Z. Similarly, in the second conventional voltage drop circuit shown in Fig. 2, a power loss between the source and drain electrodes of the FET 132 arises in the same manner as the first conventional voltage drop circuit to inevitably cause the temperature rise of the IC chip and also it is impossible to carry out a simultaneous charging of the first to third capacitors C131, C132 and C S . Hence, the utilization efficiency of the capacitors, that is, the utilization efficiency of the IC chip surface area is low and it is hard to improve the integration degree. SUMMARY OF THE INVENTION It is therefore an object of the present invention to provide a voltage drop circuit with a capacitor voltage divider circuit base in view of the aforementioned problems of the prior art, which is capable of preventing a consumption power loss and heat generation and stabilizing divided output voltages and which is suitable for high integration degree of a semiconductor IC chip. In accordance with one aspect of the present invention, there is provided a voltage drop circuit to be built in a semiconductor integrated circuit chip including a first terminal for inputting an external power source voltage, a second terminal to be maintained to a reference voltage, and an internal circuit having a predetermined function, the chip being connected to the first and second terminals, the voltage drop circuit dropping the external power source voltage and applying a dropped voltage to the internal circuit, comprising: (A) a voltage divider circuit including: (a) first and second capacitors having an equal capacity; (b) two pairs of first switch elements which are inserted between first electrodes of the first and second capacitors and the first terminal and between second electrodes of the first and second capacitors and the second terminal and which are selectively conducted in response to a control signal; (c) a pair of second switch elements which are inserted between the first electrode of the first capacitor and the second electrode of the second capacitor and between the second electrode of the first capacitor and the first electrode of the second capacitor and which are selectively conducted in response to the control signal; (d) a pair of third switch elements which are inserted in series between the first electrode of the first capacitor and the first electrode of the second capacitor and are selectively conducted in response to the control signal; (e) an output terminal connected to a connection point between the pair of third switch elements and to the internal circuit; and (f) control signal generating means for generating the control signals for controlling conduction states of the first, second and third switch elements so that the first and second capacitors are alternately charged by the external power source voltage supplied from the first terminal and are alternately discharged to the internal circuit from the output terminal in a complementary manner to the charging; and (B) connection control signal generating means for generating a connection control signal for controlling the generation of the control signals in the control signal generating means in response to a comparison result between a discharge output to the internal circuit and a predetermined comparison reference voltage. In the voltage drop circuit, the connection control signal generating means includes voltage comparator means having first and second input terminals; voltage dividing means connected to the first terminal for dividing the external power source voltage to generate the comparison reference voltage; and wiring means for connecting the first input terminal of the voltage comparator means to the output terminal and the second input terminal of the voltage comparator means to the voltage dividing means. The control signals are constructed so as not to overlap in front and rear edge portions of the alternate charge and discharge periods of the first and second capacitors. The internal circuit is a large capacity dynamic random access memory including a memory cell array for inputting a precharge voltage in response to a precharge control pulse synchronizing with a front edge of a read cycle and the output terminal is connected to a sense amplifier of the dynamic random access memory. A fourth switch element is inserted between the first terminal and the output terminal and is selectively conducted in response to the precharge control pulse. The internal circuit is a circuit of a large capacity dynamic random access memory which has a heavy-load operational period operable by a relatively heavy-current and a light-load operational period operable by a relatively light-current and which is tends to vary an output voltage on the output terminal. A supplementary voltage comparator means compares the output voltage with a supplementary comparison reference voltage lower than the comparison reference voltage, and a switch element is inserted between the first terminal and the output terminal and is selectively conducted in response to an output of the supplementary voltage comparator means. In the voltage drop circuit, a frequency counter is connected to the output of the voltage comparator means so as to detect the heavy-load operational period and the light-load operational period and detects repeat frequency of a pulse output appearing on the output of the supplementary voltage comparator means, and controller controls the supplementary voltage comparator means to selectively become an operational state in response to the output of the frequency counter and to cause the output to be supplied to the switch element. The control signal generating means includes a first bistable circuit for inputting the output of the voltage comparator means at one of a pair of input terminals, a second bistable circuit which inputs a pair of outputs of the first bistable circuit at a pair of input terminals and supplies its output as the connection control signal to the voltage dividing means and a logic circuit which gives a delay to the connection control signal and executes a pulse width conversion to supply a pulse width converted signal to another of the pair of input terminals of the first bistable circuit. In the voltage drop circuit, at least one delay circuit is connected to an output of the voltage comparator means and outputs a delay connection control signal for giving a predetermined delay to the connection control signal and at least one subordinate voltage divider circuit has substantially the same construction as the voltage divider circuit, including first and second terminals connected to the corresponding first and output terminals of the voltage divider circuit, and which inputs the delay connection control signal by control signal generating means corresponding to the control signal generating means of the voltage divider circuit. In the voltage drop circuit, three delay circuits and three subordinate voltage divider circuits can be provided. In the voltage drop circuit of the present invention, the voltage of the external power source voltage supplied to the semiconductor IC chip is divided by the voltage divider circuit of the capacitors to obtain a divided output voltage to be supplied to the internal circuit, and there is no power loss. Hence, there is no temperature increase of the IC chip. Further, the charge and discharge of a plurality of capacitors constituting the voltage divider circuit can be executed at the same time and thus the utilization efficiency of the capacitors is raised. As a result, the occupying area of the capacitors on the IC chip is reduced and this contributes to improvement of the integration degree of the whole IC chip. BRIEF DESCRIPTION OF THE DRAWINGS The objects, features and advantages of the present invention will become more apparent from the consideration of the following detailed description, taken in conjunction with the accompanying drawings, in which: Fig. 1 is a block diagram of one conventional voltage drop circuit; Fig. 2 is a block diagram of another conventional voltage drop circuit; Fig. 3a is a circuit diagram showing an equivalent circuit of the conventional circuit shown in Fig. 2 and Fig. 3b is a schematic diagram showing waveforms for explaining an operation of the conventional circuit shown in Fig. 2; Fig. 4a is a block diagram of a first embodiment of a voltage drop circuit according to the present invention and Fig. 4b is a block diagram of a control signal generator used in the circuit shown in Fig. 4a; Fig. 5a is a circuit diagram showing an equivalent circuit of a voltage divider constituting a part of the circuit shown in Fig. 4a and Fig. 5b is a schematic diagram showing waveforms for explaining an operation of the circuit shown in Fig. 4a; Fig. 6a is a block diagram of a second embodiment of a voltage drop circuit according to the present invention and Fig. 6b is a schematic diagram showing waveforms for explaining an operation of the circuit shown in Fig. 6a; Fig. 7a is a block diagram of a third embodiment of a voltage drop circuit according to the present invention and Fig. 7b is a schematic diagram showing waveforms for explaining a operation of the circuit shown in Fig. 7a; Fig. 8a is a block diagram of a fourth embodiment of a voltage drop circuit according to the present invention and Fig. 8b is a schematic diagram showing waveforms for explaining an operation of the circuit shown in Fig. 8a; Fig. 9 is a block diagram of a fifth emvbodiment of a voltage drop circuit according to the present invention; Fig. 10 is a schematic diagram showing waveforms for explaining an operation of the circuit shown in Fig. 9; Fig. 11 is a block diagram of a sixth embodiment of a voltage drop circuit according to the present invention; Fig. 12 is a schematic diagram showing waveforms for explaining an operation of the circuit shown in Fig. 11; Fig. 13a is a block diagram of a seventh embodiment of a voltage drop circuit according to the present invention and Fig. 13b is a schematic diagram showing waveforms for explaining an operation of the circuit shown in Fig. 13a; and Fig. 14 is a schematic diagram showing a rough arrangement of the fifth embodiment of the voltage drop circuit shown in Fig. 6, a plurality of which are built in an IC chip for showing an applied example of the present invention. DESCRIPTION OF THE PREFERRED EMBODIMENTS The present invention will now be described in connection with its preferred embodiments with reference to the accompanying drawings, wherein like reference characters designate like or corresponding parts throughout the views and thus the repeated description thereof can be omitted for brevity. The first embodiment of a voltage drop circuit according to the present invention will now be described in connection with Figs. 4a and 4b and Figs. 5a and 5b. Fig. 4a shows the first embodiment of the voltage drop circuit and Fig. 4b shows a control signal generator used in this embodiment. Also, Fig. 5a shows an equivalent circuit of a voltage divider constituting a part of this embodiment and Fig. 5b shows waveforms for explaining the operation of this embodiment. As shown in Fig. 4a, the voltage drop circuit 1 is comprised of a voltage divider 10 and a voltage comparator 11. A divided output voltage V INT of the voltage divider 10 is applied to one input terminal (-) of the voltage comparator 11 and a voltage at a connection point between two series resistors R11 and R12 which are inserted between a power source terminal 13 of an external power source voltage V DD and an earth electrode of an earth voltage GND is applied as a comparison reference voltage V REF to another input terminal (+) of the voltage comparator 11. An output SG11 of the voltage comparator 11 as a connection control signal is supplied to the voltage divider 10. At an output terminal 12 inputting the divided output voltage V INT , the whole internal circuits of a semiconductor IC chip with a built-in voltage drop circuit 1 are connected as a load circuit Z and a stabilizing capacitor C S by a wiring capacity is connected in parallel with the load circuit Z. The voltage divider 10 includes a voltage divider section 10a and a control signal generator section 10b for generating first, second, third and fourth control pulses Φ1B or Φ1 (an inverted signal of Φ is hereinafter referred to as Φ B or Φ ), Φ1, Φ2B or Φ2 and Φ2 for switching connections of first and second capacitors C101 and C102 in response to the connection control signal SG11. The voltage divider section 10a is comprised of a first series connection circuit which is inserted between the power source voltage V DD and the earth voltage GND and includes a first FET switch S101, a first capacitor C101, a fifth FET switch S105, a second capacitor C102, and a sixth FET switch S106 connected in series; a second series connection circuit which is inserted between the power source voltage V DD and the earth voltage GND and includes a seventh FET switch S107, the second capacitor C102, a fourth FET switch S104, the first capacitor C101 and a second FET switch S102 connected in series; a third FET switch S103 for selectively connecting the connection point among the first FET switch S101, the first capacitor C101 and the fourth FET switch S104 to the output terminal 12 of the voltage divider 10; and an eight FET switch S108 for selectively connecting the connection point among the seventh FET switch S107, the second capacitor C102 and the fifth FET switch S105 to the output terminal 12. Further, the first FET switch S101 is opened or closed in response to the first control pulse Φ1B, the fifth, sixth and eighth FET switches S105, S106 and S108 are opened or closed in response to the second control pulse Φ1, the seventh FET switch S107 is opened or closed in response to the third control pulse Φ2, and the second, third and fourth FET switches S102, S103 and S104 are opened or closed in response to the fourth control pulse Φ2. In this case, it is assumed that the first and seventh FET switches S101 and S107 are turned on when the control pulses are the low level and the second, third, fourth, fifth, sixth and eighth FET switches S102, S103, S104, S105, S106 and S108 are turned on when the control pulses are the high level. As shown in Fig. 4b and Fig. 5b, the control signal generator section 10b generates the above-described control signals varying in synchronism with the leading edges of the connection control signal. That is, the control signal generator section 10b includes a flip-flop circuit 109a which inputs the connection control signal at its input terminal C and outputs a signal SG12, a delay circuit 109b which inputs the output signal SG12 and outputs a delayed signal SG13, a pair of NAND circuit 109c and NOR circuit 109e which input the output signal SG12 output from the flip-flop circuit 109a and the delayed output signal SG13 of the delay circuit 109b at their input terminals to output the first and third control signals Φ1B and Φ2B, respectively, and a pair of inverter circuits 109d and 109f, connected to the output terminals of the respective NAND circuit 109c and NOR circuit 109e, to output the second and fourth control signals Φ1 and Φ2, respectively. In this embodiment, as shown in Figs. 4a and 4b and Figs. 5a and 5b, first, when the second control pulse Φ1 is the high level, the switches S101, S105, S106 and S108 are closed or conducting and the other switches are opened or non-conducting. In this state, the first capacitor C101 is charged on a route of the power source voltage V DD -> the first FET switch S101 -> the first capacitor C101, the second capacitor C102 connecting in parallel with the stabilizing capacitor C S via the eighth FET switch S108, and its electric charge is discharged through the load circuit Z (cycle 1). Next, when the fourth control pulse Φ2 becomes the high level, the switches S107, S104, S102 and S103 are closed and the other switches are opened. In this state, the second capacitor C102 is charged on a route of the power source voltage V DD -> the seventh FET switch S107 -> the second capacitor C102 and the the electric charge of the second capacitor C102 is discharged through the closed third FET switch S103 and the load circuit Z (cycle 2). By repeating the above-described operation in response to the connection control signal SGll becoming the active level, the divided output voltage V INT can be held in a range from (1/2) · V DD + (1/2) · ΔV to (1/2) · V DD - (1/2) · ΔV with (1/2) · V DD level as the center. Also, a period when all the switches are off is determined for preventing a power loss due to a mutual short-circuit among the capacitors C101, C102 and C S . The stabilizing capacitor C S is the wiring capacity from the voltage drop circuit 1 to the load circuit Z and has a function for holding the divided output voltage V INT to a stable value even in the period when all the switches are off. However, the level of the divided output voltage V INT in this period drops below (1/2) · V DD -(1/2)· ΔV and its level is decided depending on the stabilizing capacitor C S In the above-described circuit, the first and second capacitors C101 and C102 have an equal capacity value and respective interterminal voltages V101 and V102. As shown by thick lines in cycle 1 in Fig. 5a, the switch S101 is on in response to the low level of the first control pulse Φ 1B and the switches S105, S106 and S108 are on in response to the high level of the second control pulse Φ1, as described above. And the other switches S102. S104 and S107 are off and the capacitors C101 and C102 are connected between the switches S101 and S105 and switches S105 and S106, respectively. That is. the first capacitor C101 on the power source voltage V DD side and the second capacitor C102 on the earth voltage point side are connected in series and by charging the capacitors C101 and C102 so that the terminal voltages of the capacitors C101 and C102 may be mutually equal, the divided output voltage V INT can be expressed as follows. V INT = (1/2) · V DD By applying the divided output voltage V INT to the load circuit Z, the charging is continued to the capacitor C101 from the output terminal 12 to the power source terminal 13 as it is, but the capacitor C102 from the output terminal 12 to the earth voltage GND is discharged via the load circuit Z until it becomes the desired voltage. The operation up to this is cycle 1, as described above (Fig. 5b). Next, as shown by the thick lines in cycle 2 in Fig. 5a, when the third control pulseΦ2B becomes the low level, the switch S107 is on in response to low level of the control pulse Φ2B and the switches S104, S102 and S107 are on in response to the high level of the fourth control pulse Φ2. And the other switches S101, S106 and S108 are off and the capacitors C101 and C102 are connected between the switches S104 and S102 and switches S107 and S104, respectively. That is, the capacitor C102 on the power source voltage V DD side and the capacitor C101 on the earth voltage point side are connected in series and the capacitors C101 and C102 in cycle 1 are replaced each other. Since this connection state is caused after the charge and the discharge are executed in cycle 1, the capacitor C101 is in the charged state and the capacitor C102 is in the discharged state. Hence, the relationship between the voltage V101 of the capacitor C101 and the voltage V102 of the capacitor C102 is V101 > V102 and thus the potential of the divided output voltage V INT is increased again by the capacitor C102 in the charged state. This operation is cycle 2 (Fig. 5b). In the above-described cycle 1, the control signal generator section 10b controls the second and third control pulses Φ1 and Φ2B to the high level and the first and fourth control pulses Φ1B and Φ2 to the low level and supplies these control pulses to the voltage divider section 10a, and, when the capacitor C102 is discharged to drop the level of the divided output voltage V INT below the comparison reference voltage V REF divided by the resistors R11 and R12, the connection control signal SG11 is output from the voltage comparator 11. In response to the pulses of the connection control signal, the control signal generator section 10b changes the second and third control pulses Φ1 and Φ2B to the low level and the first and fourth control pulses Φ1B and Φ2 to the high level to switch the voltage divider section 10a to cycle 2. In cycle 2, when the capacitor C101 is discharged to drop the level of the divided output voltage V INT below the comparison reference voltage V REF , the connection control signal SG11 is output again from the voltage comparator 11. In response to the pulses of the connection control signal, the control signal generator section 10b changes the second and third control pulses Φ1 and Φ2B to the high level and the first and fourth control pulses Φ1B and Φ2 to the low level to switch the voltage divider section 10a to the above-described cycle 1. As described above. cycle 1 and cycle 2 are alternately repeated every output of the connection control signal SG11. By the above-described operation, the voltage of the divided output voltage V INT can be always held in the vicinity of (1/2) · V DD represented in formula (1). When the charge and the discharge are repeated as described above in the steady state, the variation level ΔV of the voltage is decided depending on the comparison reference voltage V REF determined in consideration of the tolerance of the voltage variation of the load circuit Z. Assuming that the connection switching of the capacitors C101 and C102 is controlled at the time point when the divided output voltage V INT becomes (1/2) · V DD - ΔV/2. the relationship between the voltages of both the ends of the capacitors C101 and C102 in cycle 1 in Fig. 5b right before the connection switching are represented as follows. V101 = V102 + ΔV V102 = V102 - ΔV Hence, the interterminal voltages V101 and V102 of the capacitors C101 and C102 can be expressed as follows. V101 = (V DD /2) + (ΔV/2) V102 = (V DD /2) - (ΔV/2) Therefore, the voltage increase of the divided output voltage V INT generated by one connection switching is V101 - V102 = ΔV. Hence, in this embodiment, the voltage of the divided output voltage V INT of the voltage drop circuit 1 is changed (it is assumed that the capacitors C101 and C102 are initially charged), as apparent from the waveform view in Fig. 5b. That is, in cycle 1, the discharge to the load circuit Z is started from the voltage (1/2) · V DD + (ΔV/2) of the divided output voltage V INT and, when reaching the voltage of (1/2) · V DD - (ΔV/2). the connection change of capacitor C101 to the lower side of the capacitor C102 is executed and then the voltage increases again to (1/2) V DD + (ΔV/2) to connect to the discharge in cycle 2. Similarly, in the following, the charge and the discharge of the capacitors C101 and C102 are repeated between the voltages (1/2) · V DD + (ΔV/2) and (1/2) · V DD -(ΔV/2). Next, the power consumed in the voltage divider section 10a in this embodiment will now be described. First, in the conventional voltage drop circuit 120 described with reference to Fig. 1. when the current I D flows in the load circuit Z applied with the divided output voltage V INT , a consumption power P s f the FET 122 is obtained as follows. P S = (V DD - V INT ) · I D Hence, assuming that V INT = (1/2) · V DD , P S (1/2) · V DD · I D . On the other hand, a power P 2 consumed by the voltage divider section 10a in the above-described embodiment is calculated (it is assumed that on-resistances of the switches S101 to S108 are very small and thus can be negligible and the capacities of the capacitors are C U and C L , respectively) as follows. P 2 = (1/2) · (C U + C L ) · {(V INTO + (1/2) · ΔV) 2 - (V INTO - (1/2) · ΔV) 2 } = (C U + C L ) · V INTO · ΔV = C U · V DD · ΔV wherein V INTO = (1/2) · V DD = {C U /(C U + C L ) } · V DD . While the divided output voltage V INT drops from the voltage V INTO + (1/2) · ΔV to the voltage V INTO - (1/2) · ΔV, the electric charge Q that the voltage V DD supplies to the voltage divider section 10a is Q = C U · ΔV. Assuming that the time required for supplying this electric charge is Δt, the current I flowing from the V DD to the voltage divider section 10a is I = Q/Δt = C U · ΔV/Δt. Hence, the power that the V DD supplies to the voltage divider section 10a is V DD I = ((C U · ΔV/Δt) · V DD and the energy given during Δt second is C U · V DD · ΔV. Therefore, the consumption power P E in due consideration of the charge current of the first capacitor C101 is represented as follows. P E = C U · V DD · ΔV Since the consumption power of the voltage divider section 10a is given by P E - P 2 , P E - P 2 = C U · V DD · ΔV -C U · V DD · ΔV = 0. Hence, when the resistances of the FET switches are negligibly small, the power consumption of the voltage divider section 10a becomes substantially zero. That is. the consumption power is less than the case of the conventional voltage drop circuit 130 shown in Fig. 2. Further, the capacitors C101 and C102 in this embodiment, one in the state of the charge and the other in the state of the discharge, supply the electric charge to the load circuit Z during both the aforementioned cycle 1 and cycle 2. In the above-described conventional example, as described above, the voltage divider circuit 134 supplies the electric charge to the load circuit Z from the capacitor C S at the charge time and supplies the electric charge to the load circuit Z from the capacitors C101 and C102 only at the discharge time. Hence, as compared with this conventional example, the utilization efficiency of the capacitors in this embodiment is high. Also, in comparison with the conventional voltage divider circuit 134 requiring a large capacity of the internal capacitor C S , in this embodiment, as described above. it is sufficient to use a certain extent of the wiring capacity from the voltage drop circuit 1 to the load circuit Z. In this embodiment, although the number of the capacitors inserted in series between the external output power source terminal and the earth voltage point in the voltage divider section 10a is two as an optimum value, of course, more than two number can be possible. Further, each of these capacitors can be composed of a plurality of element capacitors connected in parallel. In the voltage drop circuit 1 of this first embodiment, when the connection switching of the capacitors is carried out at a very short period, its divided output voltage V INT is stabilized to (1/2) · V DD , as described above. Since this voltage is changed by the variation of the V DD , in case that the comparison reference voltage V REF input to the input terminal (+) of the voltage comparator 11 is constant, the shift between the stable point of the divided output voltage V INT and the comparison reference voltage V REF becomes large and thus the normal connection switching can not be carried out. Hence, the comparison reference voltage V REF in Fig. 4a is generated from the external power source voltage V DD by the resistors R11 and R12 so that the comparison reference voltage V REF may be a slightly lower voltage (ΔV) than (1/2) · V DD , and thus, even when the voltage V DD is varied, the connection switching can be performed at a suitable timing. In the voltage drop circuit shown in Fig. 4a, the setting is carried out so that the comparison reference voltage V REF may be (1/2) · V DD - ΔV and the resistance values of the resistors R11 and R12 may satisfy the following ratio. R11/R12 = 2 · V DD / {(V DD - 2) · ΔV} - 1 These resistors R11 and R12 can be composed of diodes when the stability of the voltage V DD is high. However, in general, the variation of the voltage V DD is large and thus it is convenient to use resistors which readily track the variation. Further, in order to control the current as small as possible, the resistance values of these resistors are determined to at least 1 M ohms. Fig. 6a illustrates the second embodiment of a voltage drop circuit according to the present invention and Fig. 6b shows waveforms for explaining the operation of the voltage drop circuit shown in Fig. 6a. In this embodiment, it is clear that the voltage drop circuit 3 can be readily applied to a dynamic random access memory (DRAM). As shown in Fig. 6a, the voltage drop circuit 3 is comprised of a voltage divider 30, a voltage comparator 33 for supplying a connection control signal SG31 to the voltage divider 30, an FET switch 31 which inputs the external power source voltage V DD at the source electrode and a precharge drive signal Φ P B at the gate electrode via a terminal 34, and a stabilizing capacitor C S whose one end is connected to the earth voltage point. The voltage divider 30 has the same construction as the aforementioned voltage divider 10. The comparison reference voltage V REF is applied to one input terminal (+) of the voltage comparator 33 and the divided output voltage V INT is applied to another input terminal (-) of the voltage comparator 33, the source electrode of the FET switch 31 and the other end of the stabilizing capacitor C S . In turn, when the FET switch 31 is the on-state in response to the precharge drive signal Φ P B, the divided output voltage V INT for leading the voltage V DD supplied from an external power source terminal 36 to a divided voltage output terminal 35 is supplied to the source electrode of an FET 37 whose drain electrode is connected to a sense amplifier (not shown) and which inputs a sense enable signal SEB or SE (an inverted signal of SE) at the gate electrode, via the divided voltage output terminal 35. The operation start of the sense amplifier of the DRAM is accompanied with a momentary heavy-current and thus an insufficient current supply state is often caused. Hence, the FET switch 31 is turned on by the low level of the precharge drive signal Φ P B synchronizing with the non-operational period (the period that SEB is the high level) of the sense amplifier and the external power source voltage V DD is supplied to the FET 37 as it is. Right before coming into the operational period of the sense amplifier, that is, by making the precharge drive signal Φ P B to the high level at the state that the sense enable signal SEB is the high level and the FET 37 is off, the FET switch 31 is turned off. In this state, the node of the divided output voltage V INT is isolated from the terminal 34 of the power source voltage V DD and the sense amplifier to become a floating but the voltage is held to the level of the power source voltage V DD . When the sense enable signal SEB becomes the low level. the FET 37 is conducted and the electric charge of the node of the divided output voltage V INT is supplied to the sense amplifier. With the driving of the sense amplifier, the voltage of the divided output voltage V INT gradually drops and, when reaching the level of the comparison reference voltage V REF , the control pulses Φ1 and Φ2 of the voltage divider 30 are generated in response to the connection control signal SG31 output from the voltage comparator 33. As a result, the divided output voltage V INT increases up to (1/2) · V DD + (1/2) · ΔV. At this time, since the precharge drive signal Φ P B is the high level, the FET switch 31 is still off and the divided output voltage V INT is not reached the level of the power source voltage V DD . Next, the discharge to the sense amplifier is executed and this charge and discharge are repeated until the sense enable signal SEB becomes the high level again. While the sense enable signal SEB is the high level and the precharge drive signal Φ P B is also the high level, the level of the divided output voltage V INT at the time point when the sense enable signal SEB becomes the high level is held. Then, when the precharge drive signal Φ P B becomes the low level again, the FET switch 31 is turned on and the divided output voltage V INT increases to the power source voltage V DD level. Hence, while the sense enable signal SEB is the low level, the supply voltage level to the sense amplifier is stabilized near (1/2) · V DD . As described above, only in the precharge period right before the sense enable signal SEB actuates the sense amplifier, the external power source voltage V DD is supplied to the DRAM side to accelerate the precharge operation of the memory cells. Fig. 7a illustrates the third embodiment of a voltage drop circuit according to the present invention and Fig. 7b shows waveforms for explaining the operation of the voltage drop circuit shown in Fig. 7a. In this embodiment, as shown in Fig. 7a, the voltage drop circuit 4 is comprised of a voltage divider 40 having a similar construction as the above-described voltage divider 10, a voltage comparator 43 for supplying a connection control signal SG41 to the voltage divider 40, an FET 41 which is inserted between a power source terminal 47 and an output terminal 45 and supplies a level of a voltage (1/2) · V DD to the output terminal 45, and another voltage comparator 42 for supplying a signal SG42 to the gate electrode of the FET 41. In this case, a first comparison reference voltage V REF1 is applied to one input terminal (+) of the voltage comparator 43 from a terminal 46 and a second comparison reference voltage V REFQ is applied to one input terminal (-) of the voltage comparator 42 from a terminal 44. The divided output voltage V INT output from the voltage divider 40 is applied to another input terminal (-) of the voltage comparator 43, another input terminal (+) of the voltage comparator 42 and the output terminal 45 connected thereto and also the external power source voltage V DD , under the on-state of the FET 41, is supplied to the another input terminals (-) and (+) of the voltage comparators 43 and 42 and to the output terminal 45. A load circuit Z is connected to the output terminal 45. As apparent from Fig. 7a, in this embodiment, one voltage drop circuit of the conventional embodiment shown Fig. 1 is added to the first embodiment of the voltage drop circuit shown in Fig. 4a, and the first comparison reference voltage V REF1 can be higher than the second comparison reference voltage V REF2 . In this embodiment, the FET 41 possesses a sufficient value of on-resistance so as to make the external power source voltage V DD to be (1/2) · V DD + (1/2) · ΔV. In this embodiment, the voltage divider 40 intends to dissolve the restriction to the current supply capability due to the shortening of the connection switching period with the increase of the supply current to the load circuit Z. That is, the voltage drop circuit composed of the FET 41 and the voltage comparator 42 is combined with and arranged in parallel with the voltage divider 40 so as to prevent the level drop of the divided output voltage V INT in the period of the supply current increase to the load circuit Z. In other words, by determining the first comparison reference voltage V REF1 higher than the second comparison reference voltage V REF2 , in a light-load operation mode where a light current is applied to the load circuit Z, the divided output voltage V INT is held higher than the second comparison reference voltage V REF2 to hold the signal SG42 the high level to maintain the off-state of the FET 41. That is, in this period, the output of the voltage divider 40 constitutes the divided output voltage V INT (a period of 'light-load' in Fig. 7b). On the other hand, in a heavy-load operation mode where a heavy-current is applied to the load circuit Z, the divided output voltage V INT is below the second comparison reference voltage V REF2 and the output signal SG42 of the voltage comparator 42 is the low level. Hence, the FET 41 is on and the voltage of the divided output voltage V INT on the output terminal 45 connected to the drain electrode of the FEt 41 is increased up to the level of (1/2) · V DD + (1/2) · ΔV (a period of 'heavy-load' in Fig. 7b. Fig. 8a illustrates the fourth embodiment of a voltage drop circuit according to the present invention and Fig. 8b shows waveforms for explaining the operation of the voltage drop circuit shown in Fig. 8a. In this embodiment, the voltage drop circuit 5 has substantially the same construction as the voltage drop circuit 4 shown in Fig. 7a, except that a frequency counter 55 counts down pulses of a connection control signal SG51 output from a voltage comparator 54 and an FET 52 selectively grounds an earth terminal of another voltage comparator 53 in response to an output siganl SG53 of the frequency counter 55. In this embodiment, the frequency counter 55 counts the pulses of the connection control signal SG51 output from the voltage comparator 54 and, when its period becomes shorter the a predetermined value. the frequency counter 55 outputs the signal SG53 of the high level representing the heavy-load mode. In response to this signal SG53, the FET 52 is turned on to change an output signal SG52 of the voltage comparator 53 to the low level and make an FET 51 to be on-state and to increase a voltage on an output terminal 57 to a level of (1/2) · V DD + (1/2) · ΔV. Fig. 9 illustrates the fifth embodiment of a voltage drop circuit according to the present invention and Fig. 10 shows waveforms for explaining the operation of the voltage drop circuit shown in Fig. 9. In this embodiment, as shown in Fig. 9, the voltage drop circuit 6 is comprised of a voltage divider 60 having the same construction as the above-described voltage divider 10, a voltage comparator 61, a flip-flop circuit 62 which inputs an output signal SG61 of the voltage comparator 61. a JK flip-flop circuit 63 which is subordinately connected to the flip-flop circuit 62, a clock generator 64 for supplying a clock SG64 to the JK flip-flop circuit 63, a delay circuit 65 which inputs a connection control signal SG63 of the JK flip-flop circuit 63, a and a combination circuit composed of three inverters 66a, 66b and 66c connected in series, an NAND circuit 67 and an inverter 68, which inputs an output signal of the delay circuit 65 and generates a reset signal SG65 for clearing the flip-flop circuit 62 and the JK flip-flop circuit 63. A comparison reference voltage V REF is applied to one input terminal (+) of the voltage comparator 61 from a terminal 70 and a divided output voltage V INT is applied to another input terminal (-) of the voltage comparator 61 from the voltage divider 60. The divided output voltage V INT is supplied to a load circuit Z via an output terminal 69. In Fig. 10, the divided output voltage V INT drops below the comparison reference voltage V REF . When the output signal SG61 of the voltage comparator 61 becomes the high level, the connection control signal SG63 is generated in synchronism with the leading edge of the clock SG64 generated right after the change of the signal SG61 from the low level to the high level. In case of a short charge time corresponding to the high level period of the connection control signal SG63, the current supply capability of the voltage divider 60 drops. However, in the voltage drop circuit 6, the connection switching is executed in synchronism with the leading edge of the clock SG64 and thus a connection switching of a heavy number of voltage drop circuits can be carried out at a fixed interval. In this case, the rise of the signal SG61 to the high level is stored in the RS flip-flop circuit 62 in response to its leading edge and an output signal SG62 of the flip-flop circuit 62 is input to the J terminal of the JK flip-flop circuit 63. Hence, even when the voltage comparator 61 is reset by a power source noise or the like to change the signal SG61 to the low level before the execution of the connection switching. the stored contents of the flip-flop circuit 62 is not changed and the output signal SG62 is held to the high level. Next, the output signal SG62 is stored in the JK flip-flop circuit 63 in synchronism with the leading edge of the clock SG64 and the connection control signal SG63 as its output signal is generated. By the control pulses Φ1 and Φ2 generated in response to this connection control signal SG63, the charge of the voltage divider 60 is carried out and the divided output voltage V INT is increased to (1/2) · V DD + (1/2) · ΔV. On the other hand, the connection control signal SG63 delayed in the delay circuit 65 is input to the combination circuit composed of the inverters and the NAND circuit and the combination circuit generates one shot of the reset signal SG65. The flip-flop circuit 62 and the JK flip-flop circuit 63 are reset in synchronism with the leading edge of the reset signal SG65 and as a result. the output signal SG62 and the connection control signal SG63 are changed to the low level to finish the connection control period. At the same time, the capacitors of the voltage divider 60 are come into the discharge period and the divided output voltage V INT drops. In a 16-Mb DRAM, approximately 8 voltage drop circuits 6 are built in a chip. In Fig. 14. there is shown one embodiment of an arrangement of circuits on a heavy capacity DRAM chip with a plurality of built-in voltage drop circuits 6. In Fig. 14, the divided output voltages V INT of 8 voltage drop circuits (the same as the above-described voltage drop circuit 6) 111 to 118 are connected in common to the stabilizing capacitor C S via respective wiring elements R111 to R118 and are supplied from its common connection point to the load circuit Z which is constituted by internal circuits including the heavy capacity DRAM. A plurality of wirings from the voltage drop circuits to the load circuit Z are independently provided and by their wiring resistances, the mutual influences between the voltage drop circuits can be reduced. Hence, the outputs of the voltage drop circuits with the high divided output voltages V INT can prevent the charging of the capacitors of the voltage drop circuits with the low divided output voltages V INT . The timings of the connection control signals required for the voltage drop circuits 111 to 118 can be independently generated in the respective voltage drop circuits 111 to 118 and alternatively, can be supplied in common by a pulse source equivalent to the clock generator 64 of the voltage drop circuit 6 described above. In place of providing in the respective voltage drop circuits 111 to 118, the stabilizing capacitor or capacitors C S can be arranged near the load circuit Z to prevent a reverse charge to the voltage drop circuits 111 to 118. Also, the voltage drop circuits 111 to 118 are arranged near the external power source terminals of the power source voltages V DD and the variation of the power source voltages V DD after supplied within the chip can be controlled. Fig. 11 illustrates the sixth embodiment of a voltage drop circuit according to the present invention and Fig. 12 shows waveforms for explaining the operation of the voltage drop circuit shown in Fig. 11. This embodiment is a modification of the fifth embodiment shown in Fig. 9. In this embodiment, the voltage drop circuit 8 is comprised of a voltage divider 80 having the same construction as the above-described voltage divider 10, a voltage comparator 81, a flip-flop circuit 82 which inputs an output signal SG81 of the voltage comparator 81, a combination circuit composed of a delay circuit 83, an inverter 84, a NAND circuit 85 and another inverter 86, which converts an output signal SG82 of the flip-flop circuit 82 into signals SG84 and SG85 hereinafter described, and one shot pulse generator circuit composed of three inverters 87a, 87b and 87c, a NAND circuit 88 and an inverter 89, which generates a reset signal SG86 to be sent to the flip-flop circuit 82 from an output signal SG84 of the NAND circuit 85. In this case, the inverter 86 inputs the output signal SG84 of the NAND circuit 85 and outputs a connection control signal SG85 to be supplied to the voltage divider 80. On the other hand, the comparison reference voltage V REF and the divided output voltage V INT are applied to respective input terminals (+) and (-) of the voltage comparator 81 and the divided output voltage V INT is supplied to the load circuit Z via an output terminal 90. In Fig. 12, the divided output voltage V INT drops below the comparison reference voltage V REF and, when the signal SG81 becomes the high level, in synchronism with its leading edge, the signal SG82 is output from the RS flip-flop circuit 82. Even when the voltage comparator 81 is reset by a power source noise or the like to change the signal SG81 to the low level before the execution of the connection switching, the output signal SG82 is held to the high level. The output signal SG82 of the flip-flop circuit 82 is delayed in the delay circuit 83 and the delayed signal SG82 is fed to the inverter 84. The inverter 84 outputs an delayed inverted signal SG83. The signal SG83 is successively supplied to the NAND circuit 85 and the inverter 86 and the inverter 86 outputs the connection control signal SG85 to the voltage divider 80. By the control pulses Φ1 and Φ2 generated in response to the connection control signal SG85, the connection switching is executed in the voltage divider 80 and the divided output voltage V INT is increased to (1/2) · V DD + (1/2) · ΔV. On the other hand, the signal SG84 is kept in the high level in the period determined by the delay time in the delay circuit 83 and thus the next connection switching can not be carried out until this signal Sg84 becomes the low level. Further, the signal SG84 is converted into one shot pulse signal or the reset signal SG86 in the one shot pulse generator circuit. In synchronism with the leading edge of this reset signal SG86, the flip-flop circuit 82 is reset, and the signal SG82 is changed to the low level to input the next signal SG81. Hence, by the voltage drop circuit 8, the mutual time intervals of the connection control periods can be spread more than a predetermined value and the drop of the efficiency of the voltage drop circuit can be avoided. Fig. 13a illustrates the second embodiment of a voltage drop circuit according to the present invention and Fig. 13b shows waveforms for explaining the operation of the voltage drop circuit shown in Fig. 13a. In this embodiment, as shown in Fig. 13a, the voltage drop circuit 100 includes a voltage comparator 105, a first voltage divider 101 inputting an output signal SG101 of the voltage comparator 105, first, second and third delay circuits 106, 107 and 108 which are connected in series so as to give a predetermined delay time Δt to the input signal SG101 for each, a second voltage divider 102 inputting an output signal SG102 of the first delay circuit 106, a third voltage divider 103 inputting an output signal SG103 of the second delay circuit 107, and a fourth voltage divider 104 inputting an output signal SG104 of the third delay circuit 108. The divided output voltages V INT of the four voltage dividers 101 to 104 are applied in common to one input terminal (-) of the voltage comparator 105 and the comparison reference voltage V REF is applied to another input terminal (+) of the voltage comparator 105 from a terminal 111. The divided output voltages V INT are supplied to the load circuit Z via an output terminal 110. As shown in Fig. 13b, since the connection control signal SG101 is delayed by a certain time Δt in the delay circuits 106 to 108, the switching of the charge and the discharge in the first to fourth voltage dividers 101 to 104 is executed at mutually shifted time positions. Hence, while one of the first to fourth voltage dividers 101 to 104 is in the connection switching operational state, the other voltage dividers function as stabilizing capacitor elements. In this embodiment, by determining the delay time Δt to a longer time than the required time for the above-described connection switching in one voltage drop circuit, a mutual charge among a plurality of voltage drop circuits by the divided output voltages V INT can be prevented. As apparent to those skilled in the art, two voltage drop circuits can be provided in this embodiment. As described above, according to the present invention, in the voltage drop circuit to be built in an IC chip or the like, since a capacitor with no power loss as a circuit element for a divided voltage is used, heat generation can be prevented and this circuit is suitable for a heavy scaled LSIs. Further, since these circuit wiring elements for the divided voltages function as stabilizing capacitors, these are further effective for stabilizing the divided output voltages. Moreover, in this voltage divider, since the utilization efficiency of the capacitors for the voltage dividing is raised, the occupying area of the capacitors on the IC chip can be saved to improve the integration degree. While the present invention has been described with reference to the particular illustrative embodiments, it is not to be restricted by those embodiments but only by the appended claims. It is to be appreciated that those skilled in the art can change or modify the embodiments without departing from the scope and spirit of the present invention as defined in claim 1.

Claims: A voltage drop circuit to be built in a semiconductor integrated circuit chip including a first terminal (13) for inputting an external power source voltage, a second terminal to be maintained to a reference voltage, and an internal circuit (Z) having a predetermined function, the chip being connected to the first and second terminals, the voltage drop circuit dropping the external power source voltage (V DD ) and applying a dropped voltage to the internal circuit (Z), comprising: (A) a voltage divider circuit (10) including: (a) first (C101) and second (C102) capacitors having an equal capacity; (b) two pairs of first switch elements (S101,S102,S107,S106) which are inserted between first electrodes of the first and second capacitors and the first terminal (13) and between second electrodes of the first and second capacitors (C101,C102) and the second terminal and which are selectively conducted in response to a control signal ( 1 , 2 ); (c) a pair of second switch elements (S104,S105) which are inserted between the first electrode of the first capacitor (C101) and the second electrode of the second capacitor (C102) and between the second electrode of the first capacitor (C101) and the first electrode of the second capacitor (C102) and which are selectively conducted in response to the control signal ( 1 , 2 ); (d) a pair of third switch elements (S103,S108) which are inserted in series between the first electrode of the first capacitor (C101) and the first electrode of the second capacitor (C102) and are selectively conducted in response to the control signal ( 1 , 2 ); (e) an output terminal (12) connected to a connection point between the pair of third switch elements (S103,S108) and to the internal circuit (Z); and (f) control signal generating means (10b) for generating the control signals for controlling conduction states of the first, second and third switch elements so that the first and second capacitors (C101,C102) are alternately charged by the external power source voltage supplied from the first terminal (13) and are alternately discharged to the internal circuit from the output terminal (12) in a complementary manner to the charging; and (B) connection control signal generating means (11) for generating a connection control signal (SG11) for controlling the generation of the control signals in the control signal generating means (10b) in response to a comparison result between a discharge output to the internal circuit and a predetermined comparison reference voltage. The circuit as claimed in claim 1, wherein the connection control signal generating means includes: voltage comparator means (11) having first and second input terminals; voltage dividing means (R11,R12) connected to the first terminal (13) for dividing the external power source voltage to generate the comparison reference voltage; and wiring means for connecting the first input terminal of the voltage comparator means to the output terminal (12) and the second input terminal of the voltage comparator means to the voltage dividing means (R11,R12). The circuit as claimed in claim 1, wherein the control signals ( 1 , 2 ) are constructed so as not to overlap in front and rear edge portions of the alternate charge and discharge periods of the first and second capacitors. The circuit as claimed in claim 1, wherein the internal circuit (Z) is a large capacity dynamic random access memory including a memory cell array for inputting a ( B ) precharge voltage in response to a precharge control pulse synchronizing with a front edge of a read cycle and the output terminal is connected to a sense amplifier of the dynamic random access memory, further comprising a fourth switch element (31) which is inserted between the first terminal and the output terminal and is selectively conducted in response to the precharge control pulse ( B ). The circuit as claimed in claim 2, wherein the internal circuit (Z) is a circuit of a large capacity dynamic random access memory which has a heavy-load operational period operable by a relatively heavy-current and a light-load operational period operable by a relatively light-current and which is tends to vary an output voltage on the output terminal, further comprising: a supplementary voltage comparator means (42) for comparing the output voltage with a supplementary comparison reference voltage (V Ref2 ) lower than the comparison reference voltage (V Ref1 ), and a switch element (41) which is inserted between the first terminal and the output terminal and is selectively conducted in response to an output of the supplementary voltage comparator means. The circuit as claimed in claim 5, further including: a frequency counter (55) which is connected to the output of the voltage comparator means (54) so as to detect the heavy-load operational period and the light-load operational period and detects repeat frequency of a pulse output appearing on the output of the supplementary voltage comparator means; and means for controlling the supplementary voltage comparator means (53) to selectively become an operational state in response to the output of the frequency counter and to cause the output to be supplied to the switch element. The circuit as claimed in claim 2, wherein the control signal generating means includes: a first bistable circuit (62) for inputting the output of the voltage comparator means (61) at one of a pair of input terminals; a second bistable circuit (63) which inputs a pair of outputs of the first bistable circuit at a pair of input terminals and supplies its output as the connection control signal to the voltage dividing means; and a logic circuit (65) which gives a delay to the connection control signal and executes a pulse width conversion to supply a pulse width converted signal to another of the pair of input terminals of the first bistable circuit. The circuit as claimed in claim 2, further comprising: at least one delay circuit (106) which is connected to an output of the voltage comparator means (105) and outputs a delay connection control signal for giving a predetermined delay to the connection control signal; and at least one subordinate voltage divider circuit (102,103,104) which has substantially the same construction as the voltage divider circuit (101), including first and second terminals connected to the corresponding first and output terminals of the voltage divider circuit. and which inputs the delay connection control signal by control signal generating means corresponding to the control signal generating means of the voltage divider circuit. The circuit as claimed in claim 8. wherein three delay circuits (102,103,104) and three subordinate voltage divider circuits are provided.

Inventors: TANABE AKIRA, TANABE, AKIRA, TANABE, AKIRA, C/O NEC CORPORATION
Applicants: NIPPON ELECTRIC CO, NEC CORPORATION
Application year: 1994
Main classification: H02M3
IPC codes: G11C11, G05F3, H02M3, H03H19, G05F1