Method for recovering symbol synchronism in receivers of digitally modulated signals and circuit derived therefrom

Abstract: There is described a method for recovering symbol synchronism useable in the demodulators MPSK or QAM. In accordance with the method in question there is used a VCO inserted in a PLL for sampling the symbols at the double-frequency of the symbol frequency and for generating the symbol reading clock. The algebraic sum with alternating sign of the absolute value of four consecutive samples constitutes a generic term of an error sequence which, appropriately filtered, controls the VCO. The central values of the symbols at the reading instants are obtained by interpolating the respective samples. A hangup of the PLL is revealed by the sign discordance between samples taken in pairs. In case of detection, the sequential order of the interpolated samples is changed and the symbol reading clock phase inverted. The sign of the error sequence controlling the VCO is also inverted in order to stabilise the equilibrium point of the PLL.

Description: The present invention relates to the field of numerical transmissions and specifically concerns a method for symbol synchronism recovery in digitally modulated signal receivers and the circuit derived therefrom. As is known to correctly read sequentially transmitted symbols generally present downstream of a demodulator in the form of appropriately shaped pulses, it is necessary to read the amplitude of said pulses opposite the maximum value coinciding with the value at the central point. This assumes the use of a reading clock signal having the same cadence as the symbols, whose reading edge coincides with the instant corresponding to the central value of the symbol. Local generation of a signal having the above characteristics is rather problematical, especially because of an unavoidable if small frequency error between the signal generated locally in relation to that used in transmission. It is therefore necessary to provide appropriate synchronisation making use of the information contained in or obtainable from the data received. A method widely used today for obtaining said synchronisation is that of using a phase locked loop (PLL). The same method is also used in demodulators for digitally modulated signals, MPSK, QAM, etc., for recovery of the demodulation carrier. As concerns circuit architecture of a PLL, it has long been known but it may be useful to recall that a PLL comprises essentially a voltage controlled oscillator (VCO), a phase comparator supplying a measurement of phase shift, or phase error ε, between a reference signal and a clock signal generated by the VCO, an amplifier and lastly a loop filter of the low-pass type. The loop filter extracts from the signal emerging from the phase comparator the nearly continuous component. The latter is a function V(ε) which when appropriately amplified controls in negative feedback the VCO. The PLL first links in frequency and then in phase to the reference signal, supplying at output a 'clean' clock signal. As known, the digital modulation of a carrier by a modulating signal whose amplitude varies in a pseudo-causal manner in time, as takes place for example in vocal and television etc. signals, produces a modulated signal in which the symbol synchronism information is not contained directly. It is therefore necessary to operate an appropriate symbol distortion in reception, e.g. using a 'quadrator' or rectifier in order to obtain the above reference signal for the phase comparator. Examples of symbol, or carrier, synchronism recovery circuits using a non-linear regenerator are supplied in the volume entitled 'Phaselock Techniques' by Floyd M. Gardner published by John Wiley & Sons, second edition, pages 215-249. Said volume constitutes also a basic bibliographical reference for understanding all the problems of PLLs including that of hangup at points where the phase error ε is ±nπ with odd values of n. As known, at these points, because of the periodicity 2π of the phase error function and the sinusoidal form thereof, the voltage controlling the VCO is cancelled. The equilibrium of the system is however unstable. Indeed, the slope of the phase error function is such that a shift from one of said points does not bring back the operation of the PLL therein but makes it evolve towards a stable equilibrium point, also characterised by null phase error, which is reached only after an indefinite time. In view of the above, the symbol or carrier synchronism recovery circuits using a PLL can exhibit, due to the possible hangups, extremely long acquisition times, so much so that they cannot be used, except in special cases, in time division multiple access systems (TDMA). In the majority of known methods setting out to correct the shortcomings due to hangups of a PLL, the latter are detected with the precise purpose of adding to the control voltage VCO a pulse or other transient voltage form leading more rapidly the PLL towards the correct link point. Such methods have however the serious shortcoming of slowness in leading the PLL towards the correct link point. Indeed, to switch the operation from an unstable equilibrium point to a stable equilibrium point the PLL must always recover the phase angle ε of amplitude π in a time which cannot in the best of cases be less than the time constant of the loop filter. Therefore the purpose of the present invention is to overcome the above shortcomings and indicate a symbol synchronism recovery method comprising typical operations of a PLL whose VCO is used for generating a symbol reading clock signal, and operations aimed at correcting the phase of the above clock signal, when it is erroneously (false) locked on that of the symbol synchronism signal. The method in question allows rapid recovery of the correct symbol reading clock signal at the hangup points since it acts simultaneously both on the phase of the signal generated by the VCO and on the control voltage of the VCO. The action on the voltage is aimed only at stabilising the equilibrium point at which the PLL is currently found, avoiding a broad π phase recovery. To achieve said purposes the object of the present invention is a method of recovering the symbol synchronism in the reception of digitally modulated signals consisting of sampling the symbols at the double-frequency of the symbol frequency by using a sampling signal generated by a VCO controlled by a phase shift or phase error function existing between a synchronism extracted from the symbols and a symbol reading clock signal generated locally by division by two of the signal frequency generated by the VCO. Said error function is obtained by low-pass filtering a sequence whose individual terms are in turn obtained by calculating, with symbol cadence, an algebraic sum of samples appropriately 'distorted' for two or more symbols with said distorted samples being preceded by a negative or positive sign alternating between one sample and the next. There is also verified the existence of a sign discordance between the samples belonging to sequential sample pairs in order to invert the error function sign and the reading clock signal of the symbols generated locally, as better explained in claims 1 to 7. The method which is the object of the present invention lends itself to an immediate circuit embodiment. Therefore, another object of the present invention is a symbol synchronism recovery circuit in digitally modulated signal receivers consisting of: a PLL in which the VCO generates a signal with frequency double the symbol frequency and which is used by a symbol sampling circuit to obtain a first sample sequence, a divider by two of the signal frequency generated by the VCO to obtain a symbol reading clock signal, an adder circuit, constituting the phase comparator of the PLL, which with symbol cadence performs the algebraic sum of the absolute values of two, or multiples of two, sequential samples of said first sequence preceded by a - or + sign alternating between one sample and the next to obtain a second sequence which by passing through a low pass filter supplies a signal for control of the VCO, a hangup detection circuit for the PLL which analyses the sign of the samples and generates a hangup signal for control of a circuit performing a sample interpolation, obtaining the central values of the symbols for control of a phase inverter of the symbol reading clock and of a polarity inverter of the control signal of the VCO, as better explained in claims 8 to 16. Since the symbol synchronism recovery circuit which is the object of the present invention derives from a method in which the individual operations are for the most part additions and subtractions and combinatory logical functions, it is simple to manufacture, low in cost, and capable of operating correctly even with high symbol frequencies. In the preferred embodiment of the digital technique circuit it also becomes possible to integrate the entire circuit on a single chip, minimising costs and optimising performance. The digital technique embodiment is in line with the most modern design tendencies in the framework of MPSK or QAM modulation systems. Apart from the enormous advantages which the digital technique possesses in relation to analog, it remains however penalised by the limited operating speed of the analog-to-digital converters presently available and by that of any highly integrated circuits (VLSI). This puts a limit on the maximum performance achievable in terms of speed (bit rate). Thus privileged are the circuitry solutions using a low number of samples per symbol. The circuit which is the object of the present invention uses only two samples per symbol, equivalent to the theoretical limit of the Nyquist criterion. As concerns the method and circuit which are the object of the present invention it is necessary to make clear that the above operation of algebraic addition with alternating sign is nothing but a special application of the Fourier discrete transform with the sequence formed by the absolute values of the samples, as clarified below. Comparing the circuit which is the object of the present invention with that known consisting of a non-linear regenerator followed by a PLL there can be appreciated the considerable simplification introduced in having replaced the regenerator, i.e. distorter and selective filter, and the phase comparator of the PLL with a simple adder. An application example of the Fourier discrete transform in a synchroniser is described in the article by Martin Oerder and Heinrich Meyr entitled 'Digital Filter and Square Timing Recovery' published in IEEE TRANSACTIONS ON COMMUNICATIONS, May 1988, Volume 36, no. 5, pages 605 to 612. In the example, the transform is used to estimate the phase error ε(t) possessed by a clock generated locally and used for sampling the analog signal in base band. As stated explicitly in the conclusions of the article, the application is related to the digital embodiment of a synchronism recovery circuit of the non-feedback type in which the estimate of the delay is used to control 'forward' the interpolation units for reading the symbols, not described in the article, which act on the digitised samples of the symbols (4 samples per symbol). Said samples are obtained by sampling the symbols at fixed-frequency by means of a sampling signal generated by a free oscillator. For the calculation, precise information is necessary on amplitude and phase for the symbol frequency obtained by construing a sequence x k of numerical samples of the squared modulus of the symbol vectors. This stated, the article contains a mathematical demonstration of the fact that the expression: ε and= -1/2π arg(Xm) is an estimate of said time delay ε normalised in relation to the symbol time T, where Xm is the complex coefficient at the symbol frequency calculated by applying the Fourier discrete transform to the sequence x k as shown in formula (4) on page 605. Since the synchronisation circuit which is the object of the present invention is the 'feedback' type, at least as concerns control of the VCO, calculation of the Fourier discrete transform is amenable to a simplification in relation to the method mentioned. Indeed, more than the phase error expressed by the value of the angle ε, there becomes necessary a function thereof for control of the VCO. Said function has been identified only in the imaginary part of the complex coefficient Xm whose mathematical expression is a summation of sinusoidal type functions, just as the voltage form V(ε) at the output of a phase comparator is 'sinusoidal'. The validity of the choice is amply proven by experimental tests and by the evidence which is given by the illustration of FIGS. 6 to 10 and 6' to 10'. The great advantage which derives therefrom is that for calculation only two samples per symbol are sufficient instead of the four of the mentioned method. Further purposes and advantages of the present invention are clarified in the detailed description of an embodiment thereof given below by way of nonlimiting example with reference to the annexed drawings wherein: FIG. 1 shows a block diagram of a symbol synchronism recovery circuit SINC which is the object of the present invention, FIGS. 2 and 3 show the circuits included in the blocks indicated by HU-DET and SIM-CORR respectively in FIG. 1, FIG. 4 shows the behaviour of a function used in the blocks SIGN1 and SIGN2 of FIG. 1, FIG. 5 shows the behaviour of a function Xmi(ε) controlling the VCO of FIG. 1, FIGS. 6, 7, 8, 9 and 10 show in five different temporal situations as many sample sequences reaching the inputs of the circuit SINC of FIG. 1, placed in relationship with the respective symbols, FIGS. 6', 7', 8', 9' and 10' show as many values of the function Xmi(ε) calculated by a block DFT of FIG. 1 in the five situations of FIGS. 6 to 10, FIGS. 11 and 11' show the temporal behaviour of some symbols reaching the circuit SINC and that of the related sampling and reading signals in the case of correctly recovered synchronism, and FIGS. 12 and 12' show the temporal behaviour of the same symbols as in FIGS. 11 and 11' and that of the related sampling and reading signals in the case of synchronism not correctly recovered. With reference to FIG. 1, vp(t) and vq(t) indicate two trains of pulses in base band representing the components in phase and in quadrature respectively of a vector, termed also symbol, previously used in a modulator, not shown in the figures, to digitally modulate (MPSK, QAM, etc.) a sinusoidal carrier. Hereinafter for the sake of brevity the components vp(t) and vq(t) are also called symbols where this does not cause confusion. The pulses vp(t) and vq(t) represent the input signals of the circuit SINC and reach the input of respective analog-to-digital converters A/D-CONV1 and A/D-CONV2 to be converted in numerical samples of said pulses. The latter are indicated by p(kT/2) and q(kT/2) respectively, where T is the duration of the interval of time between two symbols, 2 is the number of samples per symbol, and k is an index identifying a generic sample in a respective sequence. The converters A/D-CONV1 and A/D-CONV2 have a sampling input to which arrives a sampling signal 2f' clk , with a frequency double of that of symbol equal to 1/T, generated by a voltage-controlled oscillator VCO. The signal 2f' clk also reaches a frequency divider by two DIV2 whose output signal at the symbol frequency is indicated by f'' clk . The numerical samples p(kT/2) and q(kT/2) reach the input of two identical blocks SIGN1 and SIGN2, respectively each of which comprises an input sample sign detector, whose output is connected to a first input of a two-input multiplier at whose second input arrives the same input sample as the respective block SIGN1 or SIGN2. The realisation of said blocks is known to those skilled in the art hence for the sake of brevity the circuits included therein are not indicated in the figures. At the output of the blocks SIGN1 and SIGN2 are present respective numerical output samples ps and qs reaching the inputs of a digital adder SOM at whose output is present a sequence of samples x k . The samples x k reach the input of a shift register SHR1, timed by the signal 2f' clk whose memory capacity is four samples x k which can be taken in parallel on four outputs where they are indicated by x₀, x₁, x₂ and x₃ and correspond to samples x k-3 , x k-2 , x k-1 , x k respectively where x k is the actual sample. The four outputs of SHR1 are methodically connected to as many inputs of a block DFT indicated by -, +, -, +. The block DFT is timed by the signal f' clk coming from DIV2. At the output of the block DFT is present a sequence xmi(T) which is sent to the input of a low pass digital filter LOOPFIL, termed also loop filter, and also timed by the signal f' clk . At the output of the filter LOOPFIL is present a filtered sequence xmif (T) which is sent to a loop stabiliser block STAB controlled by a hangup detection signal AHU generated by a block HU-DET. At the output of the loop stabiliser block STAB is present a control signal xmif'(T) reaching the numerical control input of the VCO. The numerical samples p(kT/2) and q(kT/2) arrive also at the inputs of two respective shift registers SHR2 and SHR2' timed by the signal 2f' clk . The registers SHR2 and SHR2' have a memory capacity of four samples each which can be taken in parallel on as many outputs where they are indicated by p₀, p₁, p₂, p₃ and q₀, q₁, q₂, q₃ respectively. Three outputs of SHR2 corresponding to those of the samples p₀, p₁ and p₂ are connected to as many inputs of the hangup detector block HU-DET by means of the respective busses BP0, BP1 and BP2. The fourth output of SHR2 is connected to a fourth input of HU-DET by means of a single wire FP3 on which travels the most significant bit corresponding to the sign bit of the sample (p₃) present at the fourth output of SHR2. Similarly three outputs of SHR2' corresponding to those of the samples q₀, q₁, q₂ are connected to a fifth, sixth and seventh input of the block HU-DET by means of respective busses BQ0, BQ1 and BQ2. The fourth output of SHR2' is connected to an eighth input of HU-DET by means of a single wire FQ3 on which travels the most significant bit corresponding to the sign bit of the sample (q₃) present at the fourth output of SHR2'. The block HU-DET, timed by the signal f' clk coming from the block DIV2, generates the hangup detection signal AHU which it sends to the block STAB and to a synchronism correction block SIM-CORR. The busses BP0, BP1, BP2 and BQ0, BQ1, BQ2 traverse the block HU-DET and connect to as many inputs of the synchronism correction block SIM-CORR. The latter processes the samples received and generates at output numerical values Pcor and Qcor corresponding to an estimate of the central values of the components vp(t) and vq(t) at the input of the circuit SINC. The numerical values Pcor and Qcor are present on respective output busses of the synchronism correction block SIM-CORR. To said block arrives also the signal f' clk which will be appropriately treated on the basis of the logical value taken on by the hangup detection signal AHU and made available at output as the f clk associated with the values Pcor and Qcor. With reference to FIG. 2, in which the same elements as in FIG. 1 are indicated by the same symbols, in the block HU-DET can be seen four logical gates at two inputs of the EX-OR type indicated by 1, 2, 3 and 4. The inputs of the gate 1 are connected to the wires of the busses BP0 and BP1 respectively on which travel the most significant bits, corresponding to the sign bits, of the samples present there. The inputs of the gate 2 are connected to the wire FP3 and the wire of the bus BP2 respectively on which travels the most significant bit of the sample present thereon. Similarly the inputs of the gate 3 are connected to the wires of the busses BQ0 and BQ1 respectively on which travel the most significant bits, corresponding to the sign bits, of the samples present there. The inputs of the gate 4 are connected to the wire FQ3 and to the wire of the bus BQ2 respectively on which travels the most significant bit of the sample present there. The outputs of the four logical gates 1, 2, 3 and 4 are connected to as many inputs of a logical gate 5 of the OR type. The output of the gate 5 is connected to the + input of an adder 6 and to the input of a shift register SHR3, 16 bits long, and timed by the signal f' clk (FIG. 1). The output of the register SHR3 is connected to the input of the same adder 6 whose output is connected to a first input of an add circuit 7. The second input of the adder 7 is connected to the output of the same adder where there can also be taken the hangup detection signal AHU. With reference to FIG. 3, in which the same elements as in FIGS. 1 and 2 are indicated by the same symbols, there are noted the busses BP0, BP1, BP2 and BQ0, BQ1, BQ2 (FIG. 2) entering the synchronism correction block SIM-CORR. In said block can be seen a 2-input selector 8 connected to the busses BP0 and BP2, at whose selection input arrives the hangup detection signal AHU (FIG. 2). The output of the 2-input selector 8 is connected to a first input of an interpolation circuit 10, typically a 2-input digital adder, whose second input is connected to the bus BP1. The output of the interpolation circuit 10 is connected to the input of a memory register LTCH1 timed by the signal f' clk (FIG. 1). The output of the memory register LTCH1 corresponds to the output bus of the synchronism correction block SIM-CORR carrying the numerical value Pcor. In the synchronism correction block SIM-CORR there can also be noted a 2-input selector 9 connected to the busses BQ0 and BQ2 at whose selection input arrives the hangup detection signal AHU (FIG. 2). The output of the 2-input selector 9 is connected to a first input of the interpolation circuit 11, typically a 2-input digital adder, whose second input is connected to the bus BQ1. The output of the interpolation circuit 11 is connected to the input of a memory register LTCH2 timed by the signal f' clk (FIG. 1). The output of the memory register LTCH2 corresponds to the output bus of the synchronism correction block SIM-CORR carrying the numerical value Qcor. The signal f' clk entering the synchronism correction block SIM-CORR reaches a first input of a selector 13 and an inverter 12 at whose output is present a negated signal f' clk . The output of the inverter 12 is connected to a second input of the selector 13 at whose selection input arrives the hangup detection signal AHU and at whose output is present the signal f clk representing the correct symbol reading clock signal. Operation of the symbol synchronism recovery circuit of FIGS. 1, 2 and 3 is now discussed with the aid of the remaining figures. It is premised that the input pulses vp(t) and vq(t) come from a demodulator and are previously filtered to limit their band to the value 1/T for the purposes of the subsequent sampling. The wave form of vp(t) and vq(t) is symmetrical in relation to a central value. The amplitude values can be positive, negative or null. In the case where vp(t) and vq(t) come from modulated signals MPSK transmitted with mean unit power, the amplitudes in the central points take on the values ± 1. In the case of modulation QAM said amplitudes can also take on discrete values between 0 and 1. The samples p(kT/2) and q(kT/2) at the inputs of the respective blocks SIGN1 and SIGN2 have a cadence T/2, the corresponding samples ps(kT/2) and qs(kT/2) output from said blocks take on the following expressions: ps(kT/2) = p(kT/2) × sign(x), with x = p(kT/2) qs(kT/2) = q(kT/2) × sign(x), with x = q(kT/2), where sign(x) is the transfer function of the sign detection circuit of the input samples, not shown in the figures, included in the blocks SIGN1 and SIGN2. The function sign(x) is illustrated in FIG. 4. From the foregoing it can be seen that the distortion introduced on the input sample amplitude from the blocks SIGN1 and SIGN2 consists of calculating the absolute value of said samples. At the output of the block SOM is present the sequence x k = ps(kT/2) + qs(kT/2) whose terms sum the amplitude distortion of the samples of both the symbol components and hence of the symbols themselves. The frequency spectrum of the sequence x k also contains, as known, a component at the symbol frequency signal f' clk = 1/2πT. It is also premised that the circuit of FIG. 1 comprises a digital PLL whose membership blocks are specified below. The control voltage of the block VCO is found by calculating in an appropriate manner only the imaginary part of the complex coefficient at the symbol frequency of the Fourier discrete transform of the sequence x k . Said calculation leads to a discrete function of the phase error ε whose behaviour, after filtering by the filter LOOPFIL and conversion to analog, is that of a function Xmi(ε) set forth in FIG. 5. The mathematical expression of said coefficient is a summation of products of the x k (exp-j2πk/N) type in which the imaginary part is capable of approximating the behaviour of the function Xmi(ε). The calculation of the imaginary part must be conducted considering a number of samples per symbol greater than two in order to avoid its being constantly null. By choosing the value four the calculation is reduced to a simple summation in which each addend is a sample x k multiplied by ±1 or by ±j, where j is a symbol indicating a complex value. The terms entering into said discrete function controlling the VCO are only those which in the 4-sample per symbol formula are preceded by ±j. In conclusion, the samples x k per symbol actually required are still two and correspond to physical samples coming from the sampling. The remaining two samples per symbol are fictitious samples with null value introduced for the purposes of the calculation only. The above calculation is performed by the block DFT consisting basically of a 4-input adder in which the addend present at a first input is taken with the minus sign, that present at a second input is taken with the plus sign, that present at a third input is taken with the minus sign, and finally that present at a fourth input is taken with the plus sign. Those skilled in the art are capable in the absence of further information to obtain a similar adder. The expression calculated by the block DFT during a time T is at the end as follows: sequence xmi(T) = -x₀ +x₁ -x₂ +x₃ which in more general terms becomes: xmi(T) = -x k-3 +x k-2 -x k-1 +x k . Operationally, the samples x k are memorised in SHR1 on the rising edges of the signal 2f' clk , Considering steady state operation, at each rising edge the oldest sample (x₀) is lost, the remaining samples translate by one position, and a new sample (x₃) enters into SHR1. The adder DFT is enabled on the falling edge of the signal f' clk . The calculation is finished in a time less that T/2, e.g. before the configuration of the inputs is changed. It follows that the summation is performed at symbol cadence and the same pair of samples enters therein twice. The sequence xmi(T) is designed for control of the VCO of FIG. 1. It comprises a nearly continuous component Xmi(ε) which is a function of the phase error ε existing between the symbol synchronism present in the sample sequence x k and the clock signal f' clk . In addition even the sign of Xmi(ε) agrees with that required for a control of the VCO in negative feedback. The band width of the low pass filter LOOPFIL is chosen with a compromise criterion between two opposing requirements of which a first is that of acquiring rapidly the synchronism and a second is that of having a low phase jitter in the clock signal generated locally. As known, in a PLL, synchronism acquisition time is inversely proportionate to the loop band width which coincides generally with that of the loop filter. It is also known that the phase jitter due to noise grows with increasing loop band. Therefore, desiring to widen the band of LOOPFIL to allow rapid synchronism acquisition there is unavoidably produced an increase of said phase jitter. As regards the choice of the optimal band width for the LOOPFIL of the example, it has been found that a good compromise is a FIR type 32-section digital filter. The above considerations allow identification in the following blocks: A/D-CONV1, A/D-CONV2, SIGN1, SIGN2, SOM, SHR1, DFT, LOOPFIL, DIV2, VCO and in their interconnection, the configuration of a phase loop link (PLL) provided digitally, including only partially a symbol synchronism regenerator limited to the distorter represented by the blocks SIGN1, SIGN2 and SOM. As regards the block STAB it also belongs to the PLL, but it is preferable to posticipate the function after discussing the operation of the blocks SHR2, SHR2', HU-DET and SIM-CORR. As regards operation of the above said PLL, it is now discussed with the aid of FIG. 5, FIGS. 6 to 10 and FIGS. 6' to 10'. FIG. 5 charts the above function Xmi(ε) referred to the sequence xmi(T) calculated by the block DFT of FIG. 1. The behaviour of the function Xmi(ε) corresponds to that of the nearly continuous component of the voltage emerging from a know phase comparator where the reference signal for the comparison is found through an amplitude distortion of the symbols, or samples, consisting of calculating their absolute value. This behaviour corresponds to the triangular function. Therefore, the block DFT is assimilable with a phase comparator. With the aid of FIG. 5 there are now discussed the different locking cases of the PLL of FIG. 1. As may be seen, the function Xmi(ε) has a periodicity 2π and is cancelled at points 0 and ±nπ. The convergent arrows indicate on the curve the link points of the PLL stable equilibrium operation, these points correspond to even values of n (0 included). While the divergent arrows indicate the link points in unstable equilibrium operation, otherwise termed 'hangup', corresponding to odd values of n. At the stable equilibrium points the slope of the curve is positive, contrary to what occurs at the unstable equilibrium points. The different locking situations are identified by the crossing of two lines parallel to the axis ε, of equations Xmi(ε) = ±Va, with the curve. With each of the three crossing points correspond two values of ε indicated by ±a, π±a, -π±a, respectively representing offset values of the phase of the signal f' clk compared with the symbol synchronism evaluated at the corresponding locking point. Even the sign of the function Xmi(ε) is that required for negative feedback control of a VCO. Indeed, increasing positive values of Xmi(ε), or decreasing negative values, cause increases or decreases respectively of the frequency generated by the VCO tending in both cases to annul the phase error ε. FIGS. 6 to 10 and 6' to 10' show the manner in which the function Xmi(ε) is constructed starting from the sample sequence x k . The value of the function is taken only at the more significant points. However FIGS. 6' to 10' show the behaviour also at the intermediate points. As the first step it is necessary to calculate the phase error ε given by the difference between the clock signal phase taken from the symbols and the reading clock phase thereof generated locally. The clock phase taken from the symbols and evaluated on the time axis t corresponds to T/2 (± multiples of T), while the phase of the reading clock, generated locally, is read opposite the value of t in the centre of two samples of a generic pair x₀, x₁ or x₁, x₂. It is advisable to relate the calculations to a single symbol and to the sample pair representing it in the correct locking situation. The reading clock in question is f' clk because the situation shown in FIGS. 6 to 10 anticipates any correction done on the symbols and on the clock f' clk itself. Accordingly, for each of the FIGS. 6' to 10' is shown on the abscissa the phase error ε found with the above method, the respective ordinate value is found by calculating the algebraic sum Xmi = -x0 +x1 -x2 +x3, and the four values of x are taken from the corresponding FIGS. 6 to 10. In view of the above, FIG. 6 shows a correct locking situation in which the samples x₀ and x₁ (FIG. 1) fall at time T/4 and 3T/4 from the beginning of a first symbol s j-1 and the samples x₂ and x₃ (FIG. 1) fall at times T/4 and 3T/4 from the beginning of a second symbol Sj. Calculating the phase error with reference to the symbol s j-1 and to the pair of samples x₀ and x₁ there is found ε = 0. The corresponding value of Xmi(ε) is null. FIG. 7 shows a hangup situation in which the sample x₀ falls at a time -T/4 from the end of a first symbol s j-2 , the samples x₁ and x₂ fall at times T/4 and 3T/4 respectively from the beginning of a second symbol s j-1 , and the sample x₃ falls at time T/4 from the beginning of a third symbol Sj. Calculating the phase error as in FIG. 6 there is found ε = π. In this case also the corresponding value of Xmi(ε) is null. FIG. 8 shows a situation in which the phase error is +π/2. In this case the samples x₀ and x₁ fall at times 0 and T/2 from the beginning of a first symbol S j-1 and the samples x₂ and x₃ fall at times 0 and T/2 from the beginning of a second symbol Sj. Calculating the phase error as in FIG. 6 there is found ε = +π/2. The corresponding value of Xmi(ε) is given by (+x₁) + (+x₃). FIG. 9 shows a hangup situation in which the sample x₀ falls at a time 3T/4 from the beginning of a first symbol S j-1 , the samples x₁ and x₂ fall at times T/4 and 3T/4 from the beginning of a second symbol Sj and the sample x₃ falls at time T/4 from the beginning of a third symbol S j+1 . Calculating the phase error as in FIG. 6 there is found ε = -π. In this case also the corresponding value of Xmi(ε) is null. Lastly, FIG. 10 shows a situation in which the samples x₀ and x₁ fall at times T/2 and T from the beginning of a first symbol S j-1 and the samples x₂ and x₃ fall at times T/2 and T from the beginning of a second symbol Sj. Calculating the phase error as in FIG. 6 there is found ε = -π/2. The corresponding value of Xmi(ε) is given by (-x₀) + (-x₂). FIGS. 11 and 12 show the temporal behaviour of the three symbols S j-2 , S j-1 and Sj of FIGS. 6 to 10 and the respective samples of the sequence p(kT/2) indicated in FIG. 1 by p₀, p₁, p₂ and p₃. Specifically, FIGS. 11 and 11' show a correct locking situation comparable to that shown in FIGS. 6 and 6'. As may be seen, the symbol reading instants, corresponding to the rising edges of the signal f' clk , fall exactly at the midpont thereof. FIGS. 12 and 12' show a hangup situation comparable to that shown in FIGS. 7 and 7'. In this case the symbol reading instants fall at points spaced T/2 away from the midpont thereof. There is now explained the operation of the hangup detection block HU-DET (FIG. 2) and synchronism correction block SIM-CORR (FIG. 3) which, strictly speaking, cannot be considered part of the PLL in the same way as the remaining blocks. This operation, for the sake of simplicity, refers to FIGS. 11, 11', 12 and 12' relative to only the symbols vp(t) and analogous considerations apply also for the symbols vq(t) not represented. The sequences vp(t) and vq(t) must include phase transitions allowing recovery, by means of sample amplitude distortion, of the line at the symbol frequency to be applied to the reference input of the phase comparator to allow the PLL to reach and maintain the locking. In the absence of transitions the control voltage of the VCO would be without information. In the synchronism acquisition phase an appropriate preamble sequence added during transmission at the head of the sequence of symbols to be transmitted fulfils this purpose. In the preamble sequence both components of a symbol change sign are passing from one symbol to the next. FIGS. 11 and 12 show the case of only the component vp(t). Hereinafter it is the line code of the transmitted data (scrambling) preventing generation of symbol strings with sequences of 1 or 0 too long. This stated, and in the case where vp(t) and vq(t) respect the preamble sequence, detection of a hangup consists of verifying for each sample pair p(kT/2) or q(kT/2) acquired the existence of a sign discordance between the samples, an affirmation clarified below. In practice, verification on the samples of only one of the two sequences would suffice but the analysis of both the sequences ensures greater safety. The samples p₀, p₁, p₂, p₃ and q₀, q₁, q₂, q₃ (FIG. 1) represent the status of the shift registers SHR2 and SHR2' at a particular sampling instant. Said samples are memorised on the rising edges of the signal 2f' clk as indicated in FIGS. 11' and 12'. At each updating of the registers SHR2 and SHR2' the voltage at the output of the gate 5 of FIG. 2, corresponding to a signal AHU', takes an true logical value when a discordance of values appears at the inputs of at least one of the gates 1, 2, 3 and 4, and false otherwise. Since the signal AHU' can vary with every new sample acquired, it supplies the outcome of the comparison of the sign of two consecutive samples and not of that of samples belonging to consecutive pairs. It is therefore necessary to consider the value of signal AHU' every two samples compared. This is done by the shift register SHR3 where the bit corresponding to the logical value of AHU' is loaded on the falling edge of the signal f' clk . Said register, together with the adders 6 and 7, creates a persistence on the signal AHU' obtained by comparison thereof with a threshold having hysteresis. The comparison supplies the two logical values of the hangup detection signal AHU without false detections due to disturbances. The two operations of persistence and comparison are discussed below starting from an initial situation in which the shift register SHR3 is empty and the output of the adder 7 is at value 0, corresponding to a correct link situation. Every time the signal AHU' is true, the content of SHR3 translates by one position and a 1 is loaded. If the condition of true AHU' persists for another 15 times, the output of the adder 6 is always at value 1 and the output of the adder 7 is increased up to value 16. Reaching the value 16 corresponds to the threshold for switching of the hangup detection signal AHU from false to true, indicating the occurrence of a hangup. If the previous condition on AHU' is prolonged beyond the 16 times the output of the adder 6 is zero, the output of the adder 7 persists at value 16, and the signal AHU remains unchanged. If at this point the signal AHU' becomes false, a zero reaches the + input of the adder 6 and is loaded in SHR3, still full. The output of the adder 6 takes on the value -1 and the adder 7 is decreased by one unit. If the condition AHU' false persists for another 15 times, the output of the adder 6 is still at the value -1 and the output of the adder 7 is decreased to value 0. Reaching value 0 corresponds to the threshold for switching the signal AHU from true to false, indicating thus a correct link of the PLL. If the condition AHU' is prolonged beyond 16 times, the output of the adder 6 is still zero as is the output of the adder 7 and the value of AHU remains unchanged. In view of the foregoing, the amplitude of the hysteresis mesh is 16 samples and the two thresholds correspond to the hexadecimal values 00 and 10. It now remains to clarify further the above affirmation according to which the existence of a hangup is detected by verifying for each pair of samples p₀, p₁, etc. acquired the existence of a sign discordance between the samples. In this case the clarification is supplied directly by FIGS. 12 and 12' making clear this hangup situation. As may be seen, in corrispondence of opposite falling edges of the signal f' clk the samples p₀ and p₁ compared with each other have, just as p₂ and p₃, a contrary sign because they belong to different symbols with discord sign (S j-2 , S j-1 , S j ). FIGS. 11 and 11' make clear the contrary case, i.e. that in a correct link situation said sign discordance does not exist and indeed each pair of samples belongs to a respective symbol (S j-1 , S j ). The above affirmations are based on a phase relationship between the signals 2f' clk and f' clk like that shown in the figures, corresponding to the phase relationship obtained at the output of the divider DIV2 of FIG. 1. The signal AHU is used by the loop stabiliser block STAB of FIG. 1 to stabilise the equilibrium point of the PLL in a hangup case. For this purpose, the block STAB includes a 2-input selector, not shown in the figures, at which arrive the samples of the sequence xmif(T) emerging from the loop filter LOOPFIL (FIG. 1) and the same samples with inverted sign. At the selection input arrives the signal AHU. In case of correct link of the PLL the signal AHU selects at output the samples with their actual sign, while in case of hangup the samples with inverted sign pass. The sign inversion makes positive the slope of the function Xmi(ε) at the hangup point of the PLL, i.e. it is sufficient to stabilise the equilibrium at said point, but it does not suffice to supply a correct reading clock for the symbols. Indeed, as may be seen from FIGS. 12 and 12' showing a hangup situation, the rising edge of the signal f' clk falls half way of the distance between two samples belonging to different symbols and not the same symbol as it should. The synchronism correction block SIM-CORR of FIG. 3 corrects the above by inverting the phase of signal f' clk in the case of hangup. Therefore, it supplies a reading clock f clk such that whether correct link or hangup the rising edges of the signal f clk fall half way of the distance between two samples belonging to a same symbol. Said edges thus correspond perfectly with the central values of the symbols as may be seen in FIGS. 11, 11', 12 and 12'. Therefore the signal f clk is the synchronism actually recovered that can be used for reading the symbols vp(t) and vq(t). In the circuit of FIG. 1 said reading is avoided because the synchronism correction block SIM-CORR supplies at output the values Pcor and Qcor which, unless there is a scale factor, represent the central values of the pulses vp(t) and vq(t). The value Pcor is found by adding with symbol cadence the two samples of a same symbol together as takes place for the value Qcor. The samples to be added are present in the registers SHR2 and SHR2' and have a relationship with the symbols as shown on FIGS. 11 and 12 for only the register SHR2. In view of the above, the signal AHU is suited to discriminate between the different situations, i.e. it discriminates whether two samples belonging to an actual pair of samples also belong to the same symbol. Operationally the interpolation is done by choosing a first sample, e.g. p₁, of an actual pair present in SHR2 and the choice of the second sample to be added to the first depends on the presence or absence or better on the logical value of the signal AHU. Indeed, with the signal AHU absent we have the case of FIG. 11 and the second sample is p₀. While with hangup detection signal AHU present we have the case of FIG. 12 and the second sample is p₂. Similar considerations apply in the case where as first sample there is chosen p₀ and in this case the register SHR2 must contain the symbols S j-2 and S j-1 . More generally, the above initial sample is included between two other samples which are two sampling periods apart. From the foregoing it can be seen that the interpolation involves at most three samples p(kT/2) of the four present in SHR2. At each symbol time, the above three samples, related each time to different sample pairs, are always at the same outputs of the register SHR2. Operationally, the choice of the second sample to be sent to the adder 10 is made by the logical value of the hangup detection signal AHU applied to the selection input of the 2-input selector 8 at whose signal inputs arrive the samples present on the busses BP0 and BP2 which, seen at symbol frequency, are the samples p₀ and p₂ respectively. The bus BP1 carries towards the adder 10 the above said first sample to be added which, at symbol cadence, is precisely the sample p₁. The output of the adder 10 is sampled at the signal f' clk frequency, using for the purpose the register LTCH1. Considerations similar to those for the interpolation of the samples p(kT/2) apply also for the samples q(kT/2) related to the registers SHR2' and LTCH2, to the busses BQ0, BQ1 and BQ2, to the 2-input selector 9 and to the adder 11. With reference to the circuits of FIG. 3, it is necessary to state that by the term interpolation is meant an arithmetical mean of the values of the two samples reaching the adders 10 and 11 respectively. For this purpose, the values Pcor and Qcor are found by taking the numerical values at the output of said adders and excluding from both the least significant bit, which is equivalent to dividing by two. When the phase error is null the averaged samples are equal to each other and in this case the interpolation might seem superfluous. The usefulness of the interpolation appears more evident in the presence of phase errors, in this case the two samples can differ considerably from each other so that neither of them represents a good approximation of the actual value of the symbol but this does not occur for the averaged value. The considerations concerning the non-limiting example described apply also in the case where the symbols do not have components. By assimilating in this case the symbols only with the component vp(t) all the figures remain valid provided everything related to the component vq(t) including the adder SOM of FIG. 1 is cancelled from FIGS. 1-3.

Claims: Method for recovering symbol synchronism in the reception of digitally modulated signals, the symbols, or optionally their vectorial components 'in phase' and 'in quadrature', being in the form of pulse sequences with different polarities and frequent transitions between polarities with opposite sign, characterised in that it comprises the following operations: sampling of said sequential pulses (vp(t)) with a sampling signal (2f' clk ) having a frequency double that of the symbol frequency, generated by a controlled oscillator (VCO), obtaining the samples of said symbols, division by two of the frequency of said sampling signal (2f' clk ) for generation of a reading clock signal for said symbols (f clk ), distortion of the amplitude of said samples to find a sequence of distorted samples (x k ) whose frequency spectrum comprises a symbol frequency component, calculation with symbol cadence of the phase shift, or phase error (ε), between said symbol frequency component and said symbol reading clock signal (f clk ), obtaining a control sequence (xmi(T)) of said controlled oscillator (VCO), low pass filtering of said control sequence (xmi(T)) and extraction of a nearly continuous component (Xmi(ε)) controlling said controlled oscillator (VCO) causing variations in the frequency generated tending to annul said phase error (ε), verification with symbol cadence of the existence of a sign discordance between said samples (p₀,p₁;p₂,p₃) of at least one actual sample pair (p₀,p₁), and inversion in the case of sign discordance between samples of said actual pair of the phase of said symbol reading clock signal (f clk ) and of the polarity of said nearly continuous component (Xmi(ε)). Method for recovering symbol synchronism in accordance with claim 1, characterised in that said control sequence (xmi(T)) is found by performing with symbol cadence an algebraic sum of two (x₀,x₁) or multiples of two (x₀,x₁,x₂,x₃) samples belonging to said distorted sample sequence (x k ) preceded by a negative or positive sign alternating between one sample and the next. Method for recovering symbol synchronism in accordance with claim 1, characterised in that said distorted sample sequence (x k ) is generated by calculating the absolute value of said samples (p(kT/2)) of the symbols, and in that in the case where said symbols have an 'in phase' component (vp(t)) and an 'in quadrature' component (vq(t)) said distorted sample sequence (x k ) is generated by adding with sampling cadence the absolute value (ps) of a sample (p(kT/2)) belonging to said 'in phase' component with the absolute value (qs) of a sample (q(kT/2)) belonging to said 'in quadrature' component. Method for recovering symbol synchronism in accordance with claim 1, characterised in that in the case where said symbols have an 'in phase' component (vp(t)) and an 'in quadrature' component (vq(t)) said verification of the existence of a sign discordance between said samples is done on actual pairs of samples (p₀,p₁;q₀,q₁) belonging to the one or the other of said components. Method for recovering symbol synchronism in accordance with claim 1 or 4, characterised in that said verification of the existence of a sign discordance comprises also the following operations: setting of a null initial value for the result of a repetitive counting cycle, with said initial null value constituting a first threshold of comparison of the value of said counting reaching which said sign discordance is not considered verified, increase by one unit of the value of said counting cycle upon each occurrence of a sign discordance between said compared samples (p₀,p₁;q₀,q₁), or as an alternative decrease by one unit of the value of said counting cycle upon each occurrence of a sign concordance, execution, after each said increase or decrease, of a first comparison of the result of the counting with a second threshold higher than the first and reaching which said sign discordance is considered verified, in case of increase tending to exceed said second threshold the count value being kept steadily equal to that of said second threshold, and execution, of a second comparison of said result of the count with said first threshold reaching which said discordance condition is inverted in a non discordance condition, in case of decrease tending to exceed said first threshold the count value being held steadily equal to that of said first threshold, the distance between the two thresholds being such as to create a persistence on said sign discordance or non-discordance conditions in order to avoid erroneous detections caused by the presence of disturbances. Method for recovering symbol synchronism in accordance with claim 1, characterised in that in absence of said sign discordance there is performed an interpolation between the samples of said actual pair of samples (p0,p1) belonging to a symbol (S j-1 ) or to an optional said component (vp(t),vq(t)) thereof finding an estimate of the central amplitude value of said symbol (S j-1 ) or component (Pcor,Qcor) at the reading instant, and in that in said case of sign discordance there is performed an interpolation between a first (p0,q0), or second (p1,q1) in order of time, sample of said actual pair and a third sample distant one sampling period preceding said first sample or following said second sample, respectively finding said estimate of the central amplitude value of said symbol (S j-1 ) or component (Pcor,Qcor) at the reading instant. Method for recovering symbol synchronism in accordance with claim 1 or 6 characterised in that said interpolation is an arithmetic mean of said samples. Symbol synchronism recovery circuit in digitally modulated signal receivers with said symbols, or optionally their vectorial 'in phase' and 'in quadrature' components, being in the form of pulse sequences with different polarities and frequent transitions between polarities of opposite signs, comprising: (a) a voltage controlled oscillator (VCO) generating a sampling signal by sampling means of said pulses, (b) means (12,13) for obtaining a symbol reading clock signal, (c) means (A/D-CONV1,2; SIGN1,2; 50M) for generating a sequence of distorted samples (x k ) of said pulses, (d) means (SHR1, DFT) for extracting a symbol frequency component from the frequency spectrum of said distorted sample sequence, (e) a phase comparator generating a control signal for the VCO function of the phase shift, or phase error (ε), existing between said symbol frequency component and said symbol reading clock signal and said control signal piloting the VCO in negative feedback tending to cancel out said phase error (ε), (f) a low pass or loop filter (LOOPFIL) letting pass towards the VCO only one nearly continuous component (Xmi(ε)) of said control signal, (g) means (HU-DET) for detection of 'hangups' at points distinguished by phase errors of ±nπ with n odds, characterised in that : said means (12,13) for obtaining the symbol reading clock signal (f' clk ) comprise a frequency divider (DIV2) dividing by two the frequency of the signal (2f' clk ) generated by the VCO, obtaining said symbol reading clock signal (f clk ), and comprise also a phase inverter (12,13) controlled by a hangup detection signal (AHU) which in case of hangup inverts the phase of said symbol reading clock signal (f' clk ). said detection means for 'hangups' comprise a circuit (1,2,3,4,5,6,7,SHR3) which with symbol cadence verifies the existence of a sign discordance between samples of at least one actual pair of samples (p0,p1) coming from said sampling means (A/D-CONV1,2), said verification circuit on the basis of a presence, or absence, of said sign discordance generating, or removing, said hangup detection signal (AHU), in that said nearly continuous component (Xmi(ε)) traverses a polarity reversing circuit (STAB) controlled by said hangup signal which lets pass unchanged towards the VCO said component, or reverses its sign in case of removal, or generation, respectively of said hangup signal. Symbol synchronism recovery circuit in accordance with claim 8, characterised in that said extraction means for a symbol frequency component and said phase comparator are provided by means of a first adder (DFT) performing with symbol cadence an algebraic sum of two (x₀,x₁) or multiples of two (x₀,x₁,x₂,x₃) sequential samples belonging to said distorted sample sequence (x k ) preceded by a negative or positive sign alternating between one sample and the next to obtain a sequence (xmi(T)) corresponding to said control signal of the VCO. Symbol synchronism recovery circuit in accordance with claim 8, characterised in that said generation means (SIGN1) of a distorted sample sequence (x k ) comprise a sign detector for samples coming from said sampling means (A/D-CONV1), said detector supplying at output a +1 value in correspondence of positive input samples (p(kT/2)) and a -1 value in correspondence of negative input samples (p(kT/2)), and also comprise a multiplier multiplying the values emerging from said sign detector for said input samples (p(kT/2)), obtaining absolute values (ps) of said input samples constituting said distorted sample sequence (x k ). Symbol synchronism recovery circuit in accordance with claim 8, characterised in that in the case where said symbols have 'in phase' components (vp(t)) and 'in quadrature' components (vq(t)) said generation means (SIGN1,SIGN2,SOM) of said distorted sample sequence (x k ) comprise: detectors of the sign of samples belonging to said components in phase or in quadrature supplying at output a +1 or -1 value in correspondence of positive or negative input samples respectively, multipliers multiplying the values emerging from said sign detectors for the values of respective input samples (p(kT/2),q(kT/2)), obtaining absolute values (ps,qs) of said input samples, and a second adder (SOM) performing with sampling cadence the sum of said absolute values (ps) of samples (p(kT/2)) belonging to said 'in phase' component with absolute values (qs) of corresponding samples (q(kT/2)) belonging to said 'in quadrature' component to obtain said distorted sample sequence (x k ). Symbol synchronism recovery circuit in accordance with claim 8, characterised in that in the case where said symbols have an 'in phase' component (vp(t)) and an 'in quadrature' component (vq(t)) said circuit verifying the existence of a sign discordance performs said verification on actual sample pair(p₀,p₁;q₀,q₁) belonging to one or the other of said components. Symbol synchronism recovery circuit in accordance with claim 8 or 12, characterised in that said hangup detection means (HU-DET) comprise also a circuit checking the persistence of said presence, or absence, of sign discordance and supplies a persistence value to a comparator with hysteresis comparing said persistence value with a first threshold reaching which there is generated said hangup detection signal (AHU) and with a second threshold, lower than the first, reaching which there is removed said hangup detection signal (AHU), the distance between said thresholds corresponding to a persistence value protecting said hangup detection means from erroneous detections caused by the presence of disturbances. Symbol synchronism recovery circuit in accordance with claim 8, characterised in that it comprises also one or more interpolation circuits (8,9,10,11,LTCH1,LTCH2) controlled by said hangup detection signal (AHU), said interpolation circuits performing in case of the presence of said hangup detection signal (AHU) an interpolation between the samples of said actual pair of samples (p₀,p₁) belonging to a symbol (S j-1 ), or an optional component (vp(t),vq(t)), and thereof obtaining an estimate of the central amplitude value of said symbol (S j-1 ), or component (Pcor,Qcor), at the reading instant, and performing in the case of absence of said hangup detection signal (AHU) an interpolation between a first sample (p₀) or second sample (p₁) in order of time, of said actual pair and a third sample, distant one sampling period preceding said first sample, or following said second sample, respectively obtaining said estimate of the central amplitude value of said symbol (S j-1 ) or component (Pcor,Qcor) at the reading instant. Symbol synchronism recovery circuit in accordance with claim 14, characterised in that said interpolation circuits comprise: a selector (8,9) at whose signal inputs arrive said first sample (p₀) belonging to said actual pair of samples (p₀,p₁) and said third sample (p₂) distant one sampling period after said second sample (p₁) and at whose selection input arrives said hangup detection signal AHU, said selector letting pass said first (p₀) or third (p₂) sample in case of absence, or presence, respectively of said hangup detection signal (AHU), a third adder (10,11) at a first input of which arrives a sample emerging from said selector (8,9) and at a second input arrives said second sample (p₁) belonging to said actual pair of samples (p₀,p₁), a memory register (LTCH1,LTCH2) which with symbol cadence samples the signal emerging from said third adder (10,11), ignoring the least significant bit, and obtaining said estimate of the central amplitude value of said symbol (S j-1 ) or component (Pcor,Qcor) at the reading instant. Symbol synchronism recovery circuit in accordance with claim 14, characterised in that said interpolation circuits comprise: a selector (8,9) at whose signal inputs arrive said second sample (p₁) belonging to said actual pair of samples (p₀,p₁) and said third sample distant one sampling period preceding said first sample (p₀), and at whose selection input arrives said hangup detection signal (AHU), said selector letting pass said second (p₁) or third sample in case of absence, or presence, respectively of said hangup detection signal (AHU), a third adder (10,11) at a first input of which arrives a sample emerging from said selector (8,9) and at a second input arrives said first sample (p₀) belonging to said actual pair of samples (p₀,p₁), a memory register (LTCH1,LTCH2) which with symbol cadence samples the signal emerging from said third adder (10,11), ignoring the least significant bit, and obtaining said estimate of the central amplitude value of said symbol (S j-1 ) or component (Pcor,Qcor) at the reading instant.

Inventors: MAZZOLA EZIO, MAZZOLA, EZIO
Applicants: SITS SOC IT TELECOM SIEMENS, SIEMENS TELECOMUNICAZIONI S.P.A.
Application year: 1994
Main classification: H04L7
IPC codes: H04L7