DIFFERENTIAL DETECTION METHOD AND DETECTOR USING MAXIMUM LIKELIHOOD ESTIMATION

Abstract: The phase of received waves is detected at every symbol period T by means of a phase detector (12) on the basis of the phase of a local signal and the detected phase is inputted to a serial circuit of delay circuits (15₁-15 N ) each of which lags a signal by a delay time T. Then signals with phases Ψ n (n=0, 1,..., N) delayed by 1 to N symbols are outputted and supplied to a metric calculating section (16). By adding the sum of a partial sequence (ΔΦ i ; i=n+1-q, n+2-q,..., n) out of the candidates (ΔΦ n ; n=0, 1, ..., N) of an N-symbol phase-difference sequence to the phase Ψ n-q detected at a point of time (n-q)T (q=1, 2, ..., N), to determine the estimated value of the phase Ψ n of the received waves. The v-th power of the absolute value of the difference µ n (q) between the estimated value and the phase of the received waves is used as the metric of the q-symbol phase-difference detection. Σ|µ n (q)| v =λ n , the sum of the metric from q=1 to q=n is used as a branch metric at a point of time nT for the candidates of the phase-difference sequence. A pass metric Λ=Σλ n are found by calculating the sum of the branch metrics from a point of time 1T to a point of time NT for all the candidates of the N-symbol phase-difference sequence, and the sequence candidates which give the minimum value is used as a decoding sequence.

Description: Technical Field The present invention relates to a differential detecting method for differentially detecting a digital signal transmitted as a phase difference sequence in a symbol interval and for obtaining a decoded sequence, in particular, to a differential detecting method and a differential detector thereof corresponding to a maximum likelihood sequence estimation technique. Related Art Phase modulated waves are conventionally demodulated by coherent detection and differential detection. In the coherent detection, the receive side reproduces a carrier wave as a reference signal, measures the phase of a received wave corresponding to the reference signal, and estimates a transmitted code. In this case, since the absolute phase is unknown, the sender side generally uses differential phase shift-keying modulation (DPSK) that modulates information corresponding to the variation of the phase of the carrier wave. Since the reproduced reference signal is not affected by noise and the like, low error rate can be accomplished. On the other hand, as the differential detection, differential phase detection and quadrature differential detection have been widely used. In the differential detection, the reference wave is formed of a received wave with a delay of one symbol interval. Thus, since no carrier wave reproducing circuit is required, the detecting circuit can be simply constructed and the detecting operation can be performed at high speed. Consequently, the differential detection is suitable for receiving a burst signal in time division multiple access (TDMA) communication. However, since the signal with a delay of one symbol interval is used as the reference signal, the reference signal tends to be adversely affected by thermal noise and the like. Thus, the error rate of the differential detection degrades in comparison with that of the coherent detection. As a result, depending on whether the detecting circuit is complicated, the burst signal is received, and so forth, one of the coherent detection and the differential detection is selected. For example, for four-phase DPSK, at a bit error rate 0.1%, the difference in bit energy-to-noise rate (Eb/No) between differential detection and coherent detection is 1.8 dB. To reduce the difference, a maximum likelihood quadrature differential detection for estimating a transmitted data sequence has been proposed as in Reference 1. The quadrature differential detector is composed of delay devices and multipliers. (Reference 1: D. Divsalar and M. K. Simon, "Multiple-symbol differential detection of MPSK," IEEE Trans. Commun., vol. 38, pp. 300-308, March 1990.) In addition, a technique for recursive estimation using the Viterbi algorithm has been proposed as in Reference 2. (Reference 2: D. Makrakis and K. Feher, "Optimal noncoherent detection of PSK signals," Electronics Letters, vol. 26, pp. 398-400, March 1990.) Assuming that an N-symbol phase difference sequence A n (where n = 1, 2, ..., N) is being transmitted, the received signal is quadrature-differentially detected and a maximum likelihood sequence estimation is applied. M-phase DPSK signals that are received in an interval (n-1)T ≤ t < nT can be given in the complex representation as follows. z(t) = (2Es/T) 1/2 exp j [ n +] + w(t) where  n = {2mπ/M; m = 0, 1, ..., M-1} is a modulated phase; Es is energy per symbol; T is one symbol interval;  is a phase difference between a received wave and a locally oscillated wave of the receiver; w(t) is noise of the receiver; and Δ n =  n -  n-1 is the n-th phase difference. After z(t) is filtered, it is sampled in the symbol interval. The obtained signal sample sequence is denoted by {Z n ; n = 0, 1, ..., N}. In the technique of Reference 1, a sequence that maximizes a metric given by the following equation is selected. Λ = z N + z N-1 exp j Δ N + z N-2 exp j (Δ N + Δ N-1 ) + ... + z O exp j (Δ N + Δ N-1 + ... + Δ 1 ) 2 Equation (02) can be modified as the following equation. where Re[·] is the real part of a complex number; and ( )* is a conjugate complex. In equation (03), assuming that the upper limit of the summation with respect to q is L (< N) and the following equation is defined as a branch metric phase difference sequences are successively estimated corresponding to Viterbi algorithm with M L-1 states as the technique of Reference 2. In other words, in the maximum likelihood sequence estimation according to Reference 1, metrics of all N-sequences of (N+1) received signal samples in the symbol interval T are calculated and a sequence with the maximum metric is output. Thus, the number of times of the metric calculation becomes M N . In other words, the number of times of the calculation at each time is M N /N. On the other hand, in the maximum likelihood sequence estimation corresponding to the Viterbi decoding described in Reference 2, there are M L-1 phase difference sequences. A maximum likelihood path at each time is selected. Since each state has arriving paths from M states out of M L-1 states at the just preceding time, the number of times of the metric calculation becomes M L-1 x M = M L . Thus, the number of times of the metric calculation does not depend on the length of the transmitted symbol sequence. Consequently, the number of times of the metric calculation of Reference 2 is much reduced in comparison with that of Reference 1. However, as the modulation level M of the modulation increases, the amount of calculating process exponentially increases. US-A-5,017,883 discloses a differential detection similar to the one explained above and corresponding to the prior art portions of the independent claims. An object of the present invention is to provide a maximum likelihood decoding and differential detecting method and a differential detector thereof that provide an error rate equal to or lower than a conventional maximum likelihood sequence estimation method with lesser calculation complexity or that much lower error rate with the same calculation complexity as the conventional maximum likelihood sequence estimation method without a tradeoff of maximum transmission data rate. Disclosure of the Invention This object is achieved with a method as claimed in claims 1 and 8 and a differential phase detector as claimed in claims 9 and 12, respectively. Preferred embodiments are subject-matter of the dependent claims. Brief Description of Drawings Fig. 1 is a schematic diagram showing state transition of a differential, detecting method for explaining a differential phase detecting method according to a first embodiment of the present invention; Fig. 2 is a block diagram showing the differential phase detector according to the first embodiment; Fig. 3 is a schematic diagram showing state transition of a differential phase detecting method according to a second embodiment of the present invention; Fig. 4 is a graph showing a bit error rate in a computer simulation according to the first embodiment; Fig. 5 is a graph showing a bit error rate in a computer simulation according to the second embodiment; Fig. 6 is a schematic diagram showing state transition for explaining a differential phase detecting method according to a third embodiment of the present invention; Fig. 7 is a graph showing a bit error rate in a computer simulation according to the third embodiment; Fig. 8 is a block diagram showing a quadrature differential detector according to a fourth embodiment of the present invention; Fig. 9 is a table for comparing computation complexities corresponding to the third and fourth embodiments of the present invention and to References 1 and 2 in the case that sequence length N is L; Fig. 10 is a block diagram showing a differential phase detector according to a fifth embodiment of the present invention; and Fig. 11 is a graph showing bit error rate in a computer simulation according to the fifth embodiment. Best Modes for Carrying out the Invention First, embodiments of a differential phase detecting method according to the present invention will be described. (1) First Embodiment It is assumed that the carrier phase at time n-1 modulated by M-phase DPSK modulation method is denoted by  n-1 . The phase difference representing information to be transmitted at time n is denoted by Δ n = 2mπ/M (where m = 0, 1,..., M-1). The carrier phase  n at time n is given by  n-1 + Δ n . An N-symbol phase difference sequence {Δ n ; n = 0, 1, ..., N} is transmitted. When an M-phase DPSK signal is received in an interval (n-1)T ≤ t < nT, it can be expressed as follows. z(t) = (2Es/T) 1/2 exp j ( n +) + w(t) where  n = 2mπ/M (where m = 0, 1, ..., M-1) is a modulated carrier phase; Es is energy per symbol; T is one symbol length;  is the phase difference between the received wave and the output of the local oscillator of the receiver; w(t) is noise of the receiver; and Δ n =  n -  n-1 is the n-th phase difference of information transmitted. After the received signal z(t) is filtered and the outband noise is removed, a phase difference between the received signal z(t) and the local signal of the local oscillator is detected. The phase at time n is denoted by Ψ n . The phase Ψ n is given by the following equation. Ψ n =  n + η n +  mod 2π where η n is phase noise caused by thermal noise; mod 2π is modulo operation, in which (x+2π) mod 2π = x (where |x| ≤ π). Now, as shown in Fig. 1, it is assumed that a phase sequence of (N+1) phases Ψ 0 to Ψ N are obtained at times 0 to N, respectively. The transmitted phases of symbols transmitted from the transmitter side are expressed by the following equations.  N =  N-1 + Δ N  N-1 =  N-2 + Δ N-1 . .  N-q+1 =  N-q + Δ N-q+1 . .  1 =  0 + Δ 1 From equations (3), the following equations are satisfied. On the receiver side, the variation ΔΨ N (q) of the phases of the received signals in an interval from t = (N-q)T to NT can be given by the following equations. where Δη N (q) = η N - η N-q is the phase difference noise. At this point, the fixed phase difference  between the received signal and the local oscillator of the receiver is removed. Since η N and η N-q can be approximated by independent Gaussian noise of average value = 0 and variance = 2σ 2 , Δη N (q) becomes gaussian noise with average value = 0 and variance = 2σ 2 . When a phase error vector is defined as µ = {µ N (1), µ N (2), ..., µ N (N)} T (where T is a transposed matrix), the phase noise Δη N is expressed as follows: A joint probability density function P of µ under the condition that the phase difference sequence Δ = (Δ 1 , Δ 2 , ..., Δ N ) T is transmitted can be given by the following equation. P(µ|Δ) = 1 (2π) N/2 (detR) 1/2 exp {― µ T R -1 µ 2 } where R is N × N covariance matrix of Δη = {Δη N (1), Δη N (2), ..., Δη N (N)} T ; and det R and R -1 are determinant and inverse of a matrix R, respectively. When M phase differences take place with the same probability (normally this condition is satisfied), the maximum likelihood estimation of the transmitted phase difference sequences is performed by finding the phase difference sequence Δ' = (Δ 1 ', Δ 2 ', ..., Δ N ') T that maximizes the probability of equation (7). This is equivalent to finding the phase difference sequence Δ' that minimizes µ T R -1 µ as in equation (8). To find Δ', it is necessary to obtain R. The phase difference noise is Gaussian noise with the following properties. where <x(i)> is the statistical expression of x. Thus, R can be given by the following equation. When the standard matrix theory is applied, R -1 can be given by the following equation. When R -1 is substituted into equation (8), the maximum likelihood estimation is performed by finding a phase difference sequence with the minimum value of the following equation from M N phase difference sequence candidates as shown in Fig. 1. A computer simulation shows that even if |µ n (q)| v (where v is a real number that is 1 or greater) instead of |µ n (q)| 2 is used, a similar result can be obtained. Thus, the present embodiment is arranged to find a phase difference sequence which minimizes the following value. Fig. 2 shows a construction of a differential phase detector according to the first embodiment of the present invention. A received signal is supplied from an input terminal 11. The phase difference between the received signal and the local signal of a local oscillator 13 is detected by a phase detector 12. In the case of a transmitting device for use with present mobile communication, the received phase modulated signal is an intermediate frequency signal having a center frequency at the second intermediate frequency, and is the output signal from a limiter amplifier or an AGC amplifier. The detected output is sampled by a sampling circuit 14 in a predetermined interval (symbol interval T). The sampled signal is input to delay circuits 15 1 to 15 N each of which has a delay of the symbol interval T. Thus, phases Ψ n (where n = N, N-1, ..., 0) with delays of 0 to N symbols are input to a metric calculating portion 16. In the first embodiment, whenever a predetermined number N of phases of the received signal z(t) are received in the symbol interval T, the maximum likelihood estimation is performed for a sequence of N phase differences Δ n (where n = 1 to N). In other words, the metric calculating portion 16 calculates a path metric of phase difference sequence candidate corresponding to equation (12) and stores the path metric in a metric memory 16A. The metric calculating portion 16 repeats this process for all the candidates (namely, 4 N in the case of four-phase DPSK), obtains a phase difference sequence with the minimum path metric, and outputs the result as a decoded output to a terminal 17. In reality, the metric calculating portion 16 adds the sum of partial sequence {Δ i ; i = n+1-q, n+2-q, ..., n} of the N-symbol phase difference sequence candidate {Δ n ; n = 1, 2, ..., N} to a detected phase Ψ n-q at a time (n - q)T (where q = 1, 2, ..., n) and obtains an estimated value Ψ n ' of a received signal phase Ψ n corresponding to the following equation. Ψ n ' = Ψ n-q + Δ n + Δ n-1 +···+ Δ n-q+1 mod 2π A difference (estimated error) µ n (q) between the estimated value Ψ n ' and the received signal phase Ψ n is obtained by the following equations from q = 1 to n corresponding to equation (6). µ n (1) = Ψ n - (Ψ n-1 + Δ n ) mod 2π µ n (2) = Ψ n - (Ψ n-2 + Δ n + Δ n-1 ) mod 2π . . µ n (q) = Ψ n - (Ψ n-q + Δ n + Δ n-1 + ... + Δ n-q+1 ) mod 2π . . µ n (n) = Ψ n - (Ψ 0 + Δ n + Δ n-1 + ... + Δ 1 ) mod 2π Next, it is assumed that the v-th power value of the absolute value of each estimated error µ n (q) is a metric of q-symbol differential phase detection. The summation of the metrics from q = 1 to n at a time nT is given by the following equation. As the most important point of the present invention, the phase error at any time n is estimated not only from the phase Ψ n-1 of the just preceding signal, but also from phases Ψ n-2 , ... of further preceding signals, and thus the bit error rate can be improved using these estimated errors µ n (q). As in equation (12), v is theoretically 2. However, simulation results show that v can be any positive real number in the range from 1 to 10. By the summation of the branch metrics from time 1T to time NT, the path metric Λ of the phase difference sequence candidates Δ 1 , Δ 2 , ..., Δ N is obtained corresponding to the following equation. Path metrics are calculated for all (M N ) N-symbol phase difference sequence candidates and then stored in a metric memory 16A. A phase difference sequence candidate with the minimum path metric is defined as a decoded sequence and output from a terminal 17. (2) Second Embodiment In the first embodiment, the error rate is improved in proportion to the value of N. However, in the M-phase DPSK system, since there are M phase points, path metrics for all the M N sequence candidates shown in Fig. 1 should be calculated. Thus, the computation complexity is exponentially proportional to the value of N. In a second embodiment of the present invention, to satisfy the improvement of the error rate and the reduction of the computation complexity, the upper limit of the addition with respect to q in equation (13) is set to L (where L is any integer; 2 ≤ L < N). In addition, the following equation is defined as a branch metric. As shown in a state transition of Fig. 3, phase difference sequences are successively estimated using Viterbi algorithm with M Q-1 states. In this case, states S n at a time nT are defined by a phase difference sequence {Δ n , Δ n-1 , ..., Δ n-Q+2 } from time n-Q+2 to time nT. Since each phase difference has M values, the number of states at each time is M Q-1 . The relation between Q and L is 2 ≤ Q ≤ L. The construction of a phase difference detector according to the second embodiment will be described with reference to Fig. 2. In this embodiment, the number of delay circuits 15 connected in series is L rather than N. Next, the process algorithm performed in the metric calculating portion 16 shown in Fig. 2 will be described. Step S1: When metrics λ n (S n-1 → S n ) of M branches from M Q-1 states S n-1 = (Δ n-(Q-1) , ..., Δ n-2 , Δ n-1 ) at a time (n-1)T to a state S n = (Δ n-(Q-2 ), ..., Δ n-1 , Δ n ) at a time nT are calculated, surviving path to the state S n-1 is traced back for a (L-1) time and a sequence of (L - 1) symbols (Δ n-i ; i = 1, 2, ..., L-1) is obtained. A phase difference Δ n at a time nT that is a last symbol is added to the symbol sequence to obtain an L-symbol sequence and then the metric λ n (S n-1 → S n ) is calculated corresponding to equation (18). The number of states at each time is M Q-1 . However, the number of states S n-1 at the time (n-1)T that can be shifted to each state S n at the immediately succeeding time nT is M out of M Q-1 (where M is the number of phases of the DPSK system). Step 2: A path metric of M branches to the state S n is calculated corresponding to the following equation. Λ n (S n | S n-1 ) = Λ n-1 (S n-1 ) + λ n (S n-1 → S n ) The path with the minimum path metric is a surviving path with the maximum likelihood that arrives at the state S n , and the state from which the transition has been made is stored in the path memory 16B (see Fig. 2). The value of the minimum path metric is defined as a path metric Λ n (S n ) of each state and stored in the metric memory 16A. Step S3: A path that gives the minimum path metric Λ n (S n ) among M Q-1 states S n is selected as a surviving path and the state transition stored in the path memory 16B is traced back along the selected path for an interval DT. The depth of the trace back may be about 4. Phase difference Δ n-D of state (Δ n-(D+Q-2) , ..., Δ n-(D+1) , Δ n-D ) traced back are output as decoded results Δ n-D' . Fig. 4 shows the results of a computer simulation of the error rate characteristics of the differential phase detection according to the first embodiment. In the simulation, v = 2. In the graph, the horizontal axis represents bit energy-to-noise ratio (Eb/No). In Fig. 4, the cases of N = 2, 3, and 4 are plotted by ○, Δ, and □, respectively. In addition, for comparison, error rates of conventional symbol-by-symbol differential phase detection (N = 1) and coherent detection with differential decoding are plotted by x and +, respectively. In Fig. 4, solid lines 21 and 22 represent ideal curves. To accomplish the error rate of 0.1%, the difference in Eb/No between symbol-by-symbol differential phase detection and the coherent detecting with differential decoding is 1.8 dB. When N = 3, the difference in Eb/No can be reduced to 0.9 or less. When N = 4, the difference in Eb/No becomes around 0.6 dB. In Fig. 4, the theoretical characteristics (Reference 1) of error rate in the case that the maximum likelihood sequence estimation is performed for the conventional quadrature differentially detected output are denoted by dotted lines. From Fig. 4, it is clear that in the first embodiment, almost the same improvement as Reference 1 can be accomplished. Fig. 5 shows the results of a computer simulation of the error rate characteristics of the differential phase detection method according to the second embodiment. When L = Q = 4, the difference in Eb/No between the symbol-by-symbol differential phase detection and the coherent detection with differential decoding becomes approximately 0.2 dB. In the second embodiment, the relation between Q and L is 2 ≤ Q ≤ L. Next, a third embodiment where Q = 2 and Q < L will be described. (3) Third Embodiment In the second embodiment, the Viterbi algorithm with M Q-1 states (where 2 ≤ Q ≤ L) was used so as to reduce the computational complexity in comparison with that of the first embodiment. In the third embodiment, a differential phase detection method with lesser computational complexity will be described. As with the second embodiment, in the third embodiment, the maximum likelihood sequence estimation is performed based on the Viterbi algorithm. As shown in a state transition diagram (M = 4) of Fig. 6, in the third embodiment, the number of states of the Viterbi algorithm is M that is the same as the number of phases of modulation. In other words, in the third embodiment, Q = 2. Thus, a state Sn at a time nT is Δ n . The phase difference represents the state. Consequently, there are only M survival paths at each time. In the second embodiment, the number of survival paths at each time is M Q-1 . Next, the construction of the third embodiment will be described with reference to Fig. 2. First, a phase Ψ n of a received signal z(t) with respect to a locally oscillated signal is detected by the phase detector 12 in the transmitted symbol interval T. As described above, a detected phase at a time nT is Ψ n =  n +  + η n where η n is phase noise caused by thermal noise. (L+1) received phase samples {Ψ n-q ; q = 0, 1, ..., L} are used for the predetermined value L. As is clear from equation (5), Ψ n and Ψ n-q have the following relation. Phase errors µ n (1), µ n (2), ..., µ n (L) between the received phase Ψ n and the estimated phase Ψ n ' are calculated corresponding to the following equation. As with equation (18), the sum of the v-th power values of the absolute values of the phase errors are calculated from q = 1 to L as a branch metric λ n . Thus, the branch metric can be given by the following equation. where v is any real number that is 1 or greater. With the branch metric expressed by equation (22), symbols are decoded using Viterbi algorithm with M states (that will be described later). Step S1: To select a most likely path that arrives at a state Δ n at a time nT from M phase difference states at a time (n-1)T, a phase difference sequence {Δ n-1 ; i = 1, 2, ..., L-1) stored in the path memory 16B is read by tracing back surviving paths for a past time (n-L+1) starting from one state Δ n-1 of M states at a time (n-1)T. The state Δ n at the time nT that is a last state is added to the phase difference sequence. Thus, the phase difference sequence candidate {Δ n-i ; i = 0, 1, ..., L-1} is formed. Step S2: A detected phase Ψ n-q at a time (n-q)T is added to the sum of the phase differences of the partial sequence {Δ n-i ; i = 0, 1, ..., q-1} of the phase difference sequence candidate so as to obtain an estimated value Ψ n ' of a phase Ψ n . A phase error µ n (q) that is the difference between the estimated value Ψ n ' and the detected phase Ψ n is obtained from q = 1 to L corresponding to equation (15). Step S3: The v-th powers of the absolute values of the L phase errors µ n (q) are calculated. The resultant values are added from q = 1 to L corresponding to equation (22). Thus, a branch metric λ(Δ n-1 → Δ n ) that represents likelihood of transition from the state Δ n-1 at the time (n-1)T to the state Δ n at the time nT is obtained. Next, as in the following equation, the branch metric is added to the path metric Λ(Δ n-1 ) of the state Δ n-1 at the time (n-1)T stored in the metric memory 16A so as to obtain a path metric Λ(Δ n | Δ n-1 ) of a candidate sequence that passes through the state Δ n-1 . Λ(Δ n | Δ n-1 ) = Λ(Δ n-1 ) + λ(Δ n-1 → Δ n ) Step S4: The above-described calculations are performed for each of the M states Δ n-1 at the time (n-1)T so as to obtain path metrics for the M candidate sequences leading to the state Δ n . By comparing the path metrics, a state Δ n-1 ' with the minimum value is obtained. The state Δ n-1 ' that is decided as a state at the time (n-1)T on the most likely path to the state Δ n at the time nT is stored in the path memory 16B. The path metric Λ(Δ n | Δ n-1 ') is decided as a path metric Λ(Δ n ) of the state Δ n at the time nT and is stored in the metric memory 16A. Step S5: The processes and calculations at the steps S1 to S4 are repeated for all M states Δ n at the time nT so as to obtain M path metrics. By comparing the M path metrics, a state Δ n ' with the minimum value is obtained. The path memory is traced back for a predetermined interval DT starting from the state Δ n ' and the obtained state Δ n-D is output as a decoded symbol. Fig. 6 shows an example of a state transition diagram in the case of M = 4 according to the third embodiment. In this example, the maximum likelihood path to the state Δ n = 0 at the time nT is selected. Referring to Fig. 6, paths denoted by dotted lines extend from four states Δ n-1 = 0, π/2, π, and 3π/2 at the time (n-1)T to Δ n = 0. Survival paths denoted by solid lines extend to each state at the time (n-1)T. For example, in the calculation for the path metric including a transition branch from one state Δ n-1 = π/2 at the time (n-1)T to one state Δ n = 0 at the time nT, the state is traced back for (L-1) states along a survival path (stored in the path memory 16B) that extends to the state Δ n-1 = π/2 at the time (n-1)T. Thereafter, a branch metric λ(Δ n-1 → Δ n ) is calculated corresponding to equations (15) and (22). The branch metric is added to the path metric Λ(Δ n-1 ) of the state Δ n-1 = π/2 at the time (n-1)T stored in the metric memory 16A so as to obtain a path metric Λ(Δ n | Δ n-1 ) of the candidate sequence. Such a process is repeated for the M states Δ n-1 at the time (n-1)T so as to obtain respective path metrics. By comparing the M path metrics, the maximum likelihood path to the state Δ n at the time nT is selected. Fig. 7 is a graph showing the results of a computer simulation of error rate performances of the four-phase DPSK method according to the third embodiment. In this simulation, v = 2. In the graph, the horizontal axis represents the ratio of signal energy and noise power density per bit (Eb/No) and the vertical axis represents the error rate. For comparison, in Fig. 7, simulation results of error rates of the conventional symbol-by-symbol differential phase detection (L = 1) and the coherent detecting with differential decoding are plotted. In the graph, solid lines represent theoretical curves. The difference in Eb/No between the symbol-by-symbol differential phase detection and the coherent detecting with differential decoding at the error rate of 0.1% is 1.8 dB. However, when L = 2, the difference in Eb/No is the half or less of 1.8 dB. In Fig. 7, theoretical performance curves of the error rate of the maximum likelihood sequence estimation of Reference 1 are plotted with dotted lines. When L = 2 and 4, performances equivalent to L = 3 and 5 in Reference 1 are accomplished, respectively. As the value of L increases, the performance close to Reference 1 is accomplished. Thus, in the case of performing the most likelihood sequence estimation through the Viterbi algorithm, when the number of states of the Viterbi decoder is equal to the number of phase of the modulation, the computational complexity can be remarkably reduced in comparison with that of Reference 1. (4) Fourth Embodiment In the third embodiment, the phase difference between the received signal z(t) and the locally oscillated signal of the local oscillator 13 is detected by the phase detector 12 so as to perform the Viterbi decoding with the M states. However, as with the third embodiment, the Viterbi decoding with M states can be performed for a sample sequence of a complex detection output of a quasi-coherent detection for the received signal z(t). This method will be described as a fourth embodiment of the present invention. Fig. 8 is a block diagram showing the construction of a quadrature differential detector according to the fourth embodiment. A received signal z(t) is supplied from an input terminal 11 to a quasi-coherent detecting circuit 12. The quasi-coherent detecting circuit 12 performs quasi-coherent detection for the input signal with two locally oscillated signals supplied from a local oscillator 13. The frequency of the input signal is nearly the same as that of the locally oscillated signals. The locally oscillated signals are 90 deg. apart in phase. The complex output of the quasi-coherent detecting circuit is supplied to a sampling circuit 14. The sampling circuit 14 samples the complex signal in a predetermined interval (symbol interval T) and outputs a complex sample Z n of the received signal. The complex sample Z n is input to delay circuits 15 1 to 15 L that are connected in series and each of which causes delay of a symbol period T. The delay circuits 15 1 to 15 L output samples with delays of 1 to L symbols {Z n-q ; q = 1, 2, ..., L}. The delayed samples and non-delayed samples Z n are input to a metric calculating portion 16. The metric calculating portion 15 has a metric memory 16A and a path memory 16B. The metric calculating portion 16 performs calculations corresponding to a decoding algorithm similar to that in the third embodiment. The decoding algorithm will be described in the following. Decoded output data is obtained from a terminal 17. Step S1: To select a most likely one of the paths arriving at a state Δ n at a time nT from M phase difference states at a time (n-1)T, a surviving path stored in the path memory 16B is traced back to a past time (n-L+1) starting from a state Δ n-1 of M states at the time (n-1)T so as to read a phase difference sequence {Δ n-i ; i = 1, 2, ..., L-1}. The state Δ n at the time nT as-the last state is added to the phase difference sequence so as to form a phase difference sequence candidate {Δ n-i ; i = 0, 1, ..., L-1}. Step S2: The phase of the received signal sample Z n-q is rotated for the sum of the partial sequence (Δ n-i ; i = 0, 1, ..., q-1} of the sequence candidate. This process is repeated from q = 1 to L. The L obtained values are added so as to obtain an estimated value z n ' of the received signal sample z n . Step S3: As expressed in equation (04), a real value of the inner product of the received signal sample z n and the estimated value z n ' is defined as a branch metric λ(Δ n-1 → Δ n ) that represents the likelihood of the transition from the state Δ n-1 at the time (n-1)T to the state Δ n at the time nT. The branch metric is added to a path metric Λ(Δ n-1 ) of the state Δ n-1 at the time (n-1)T as in equation (23) so as to obtain a path metric Λ(Δ n | Δ n-1 ) of a candidate sequence that passes through the state Δ n-1 . Step S4: The above-described calculations are repeated for the M states Δ n-1 at the time (n-1)T so as to obtain path metrics for the M candidate sequences. By comparing the path metrics, a state Δ n-1 ' with the maximum value is obtained. The state Δ n-1 ' is decided as a state at the time (n-1)T of the most likely path to the state Δ n at the time nT and stored in the path memory 16B. The path metric Λ(Δ n | Δ n-1 ') is defined as a path metric Λ(Δ n ) of the state Δ n at the time nT and stored in the metric memory 16A. Step S5: The processes and calculations at the steps S1 to S4 are repeated for all M states Δ n at the time nT so as to obtain M path metrics. By comparing the M path metrics, the state Δ n ' with the maximum value is obtained. The path memory is traced back for a predetermined interval DT starting from the state Δ n ' and the obtained state Δ n-D is output as a decoded symbol. In the quadrature differential detecting based on the Viterbi decoding described in Reference 2, there are M L-1 survival paths at each time. However, in the methods according to the third and fourth embodiments of the present invention, since there are only M survival paths, the error rate thereof are slightly inferior to that of Reference 2. However, in these embodiments, the calculating amount of the branch metrics at each time is only M 2 . Thus, the computational complexity in the third and fourth embodiments is much smaller than that of Reference 2. Fig. 9 shows the number of calculations of the branch metrics at each time in the case of M = 4. According to the results of the simulation of the method proposed in Reference 2, when the Viterbi algorithm with M L-1 states is used, the error rate performance equivalent to 2L in Reference 1 can be accomplished. The number of calculations of the branch metrics for obtaining the error rate performance equivalent to the case of L = 6 in Reference 1 is compared among the methods of References 1 and 2 and the present invention. The number of branch metric calculations in Reference 1 is 683; the number of calculations in Reference 2 is 4096; and the number of calculations in the present invention is 16. Thus, it is clear that the number of calculations in the present invention can be remarkably reduced in comparison with that of the conventional methods. (5) Fifth Embodiment In the third and fourth embodiments, the Viterbi algorithm is used. Thus, one most likely path leading to each of a predetermined number of states at each time from the just preceding time is selected. Symbol corresponding to states at a time traced back for a predetermined interval along the most likely path selected among them are output as a decoded result. Thus, the decoded result has a delay against a received phase sample for the predetermined number of symbols. As a maximum likelihood decoding method that is free of such a delay, a decision feedback decoding algorithm that does not use the Viterbi algorithm is known. In the decision feedback decoding algorithm, there is only one surviving path at each time. Based on the single path of a sequence that has been determined, the next state transition is determined. The decision result is immediately output. Next, the decision feedback decoding method will be described as a fifth embodiment of the present invention. Fig. 10 is a block diagram showing the construction of a differential phase detecting circuit according to the fifth embodiment of the present invention. A phase modulated signal z(t) is received from an input terminal 11. The phase modulated signal z(t) is supplied to a phase detector 12. The phase Ψ(t) of the phase modulated signal z(t) is detected corresponding to the phase of a locally oscillated signal of a local oscillator 13. An output of the phase detector 12 is supplied to a sampling circuit 14. The sampling circuit 14 samples the input signal in a symbol interval T and outputs a phase sample Ψ n of the received signal to a differential phase detecting portion 15. In the differential phase detecting portion 15, the detected phase Ψ n of each symbol is supplied to L delay circuits 15 1 , 15 2 , ..., 15 L each of which has a delay of one symbol interval T. The differences Ψ n - Ψ n-1 , Ψ n - Ψ n-2 , ..., Ψ n - Ψ n-L between the delay outputs Ψ n-1 , Ψ n-2 , ..., Ψ n-L of the delay circuits 15 1 , 15 2 , ..., 15 L and the input detected phase Ψ n are obtained by subtracting circuits 15S 1 , 15S 2 , ..., 15S L . The phase differences accord with those expressed by equation (5). The phase differences are supplied to a metric calculating portion 16. The metric calculating portion 16 decides on the phase difference Δ  n corresponding to a calculating process as will be described later and outputs it to an output terminal 17. In addition, the phase difference Δ  n is supplied to a cumulating portion 18. In the cumulating portion 18, the detected phase difference Δ  n is input to (L-1) delay circuits 18 1 , 18 2 , ..., 18 L-1 , each of which is connected in series and has a delay of T. The delay circuits 18 1 , 18 2 , ..., 18 L-1 supply delay outputs outputs Δ  n-1 , Δ  n-2 , ..., Δ  n-L+1 to adding circuits 18A 1 , 18A 2 , 18A 3 , ..., 18A L-2 , respectively. Outputs of the adding circuits 18A 1 , 18A 2 , ..., 18A L-2 are successively supplied to next-stage adding circuits 18A 2 , 18A 3 , ..., 18A L-2 . Outputs of the adding circuits 18A 1 to 18A L-2 are supplied to the metric calculating portion 16. In other words, δ n-1 (q) = ΣΔ  n-i (where Σ is the sum from i = 1 to q and q = 1 to L-1) is supplied to the metric calculating portion 16. The metric calculating portion 16 selects one candidate from M phase differences Δ n ' (in the case of four-phase DPSK system, 0, π/2, π, and 3π/2) and adds the selected one to each added value δ n-1 (q) supplied from the cumulating portion 18. The v-th power value of the absolute value or the difference µ n (q) (corresponding to equation (6)) of the selected candidate phase difference alternative and the detected phase difference (Ψ n -Ψ q ) is obtained as follows. µ n (1) v = Ψ n - Ψ n-1 - Δ n ' v µ n (2) v = Ψ n - Ψ n-2 - (Δ n ' + Δ  n-1 ) mod 2π v µ n (3) v = Ψ n - Ψ n-3 - (Δ n ' + Δ  n-1 + Δ  n-2 ) mod 2π v µ n (L) v = Ψ n - Ψ n-L - (Δ n ' + Δ  n-1 + ... + Δ  n-L+1 ) mod 2π v where v is a real number that is 1 or greater. The sum λ n = Σ|µ n (q)| v of |µ n (l)| v to |µ n (L)| v is defined as the branch metric of the candidate phase difference Δ n '. For all M phase difference candidates Δ n ', their branch metrics are calculated and the candidate phase difference Δ n ' with the minimum branch metric is output as the detected phase difference Δ  n . Fig. 11 is a graph showing results of a computer simulation of error rate performance of four-phase DPSK system according to the fifth embodiment. In this simulation, v = 1. In the graph, the horizontal axis represents bit energy-to-noise ratio (Eb/No). In Fig. 11, the performance in the case of L = 1 is the same as that of the conventional symbol-by-symbol phase delay detection. In Fig. 11, for comparison, the error rate performance of the coherent detecting and differential decoding is shown. The difference in Eb/No between the symbol-by-symbol differential phase detection and the coherent detecting with differential decoding at the error rate of 0.1% is 1.8 dB. When L = 3, the difference in Eb/No can be reduced to almost the half of 1.8 dB. When L = 10, the difference in Eb/No becomes 0.2 dB. As described above, in the differential detection method according to the present invention, the error rate can be much more improved than that of the conventional symbol-by-symbol differential detection. Thus, the error rate performance close to the coherent detection with differential decoding can be accomplished. Alternatively, when the error rate performance attainable with conventional method is to be attained, the calculating amount can be much reduced in comparison with the conventional method. In the first, second, third, and fifth embodiments of the present invention, since estimation of maximum likelihood phase difference sequence is performed using detected phase sequences of received signal, a practical limiter amplifier can be used. In addition, since a fast acquisition property that is a feature of the differential detection is not lost, the present invention can be applied to burst reception in the TDMA system.

Claims: A differential detection method of an M-phase DPSK modulated signal, comprising the steps of: a) detecting a phase Ψ n of a received signal relative to a local signal at each time nT of a predetermined transmitted symbol interval T , where n is any integer; b) producing candidates of phase difference sequences, calculating a path metric for each candidate, and subjecting the detected phase Ψ n to a maximum likelihood sequence estimation to determine the phase based on the path metrics; characterized in that said step b) includes: b-1) tracing back along a path of each of plural N-symbol phase difference sequences {Δ n ; n=1, 2, ..., N} for q symbols so as to form a partial sequence {Δ i ; i=n, n-1,..., n+1-q}, and adding a sum of the partial sequence to a detected phase Ψ n-q symbols before so as to obtain an estimated phase value Ψ n ' of the phase Ψ n ; b-2) repeating the step b-1) for a plurality of values of q between 1 and n, inclusive, to produce a plurality of estimated phase values for said path, obtaining a phase error µ n (q) between each of the estimated phase values Ψ n ' and the phase Ψ n and defining the v-th power value of the absolute value of the phase error µ n (q) as a metric of the estimated phase value, where v is a real number that is 1 or greater; b-3) summing up the metrics for said estimated phase values to obtain a branch metric λ n for said path and summing up branch metrics along said path to obtain a path metric Λ; and b-4) defining an N-symbol phase difference sequence with the minimum path metric as a decoded sequence and outputting the decoded sequence. The differential detection method according to claim 1, wherein said step b-3) includes the steps of: b-3-1) calculating each branch metric by λ n = µ n (1) v + µ n (2) v + ... + µ n (n) v ; and b-3-2) adding the branch metric λ n from n = 1 to N so as to obtain said path metric Λ = λ 1 +λ 2 +...λ N for the path of phase difference sequences {Δ n ; n= 1,2,,..,N}. The differential detection method of claim 1, wherein the differential detection method uses a path memory and a path metric memory, the path memory being adapted for storing, as a surviving path, M Q-1 states defined by Q modulated phase differences at each time and surviving paths each representing from which one of states at the just preceding time a most likely path arrives at each of the M Q-1 states, where Q is a predetermined integer that is 2 or greater, and the path metric memory being adapted for storing the path metric that represents the likelihood of a sequence leading to each state, wherein; said step b- 1) includes the steps of: b-1-1) tracing back from one state S n-1 of the M Q-1 states at a time (n-1)T along a surviving path stored in the path memory for an (L-1) time, obtaining a sequence {Δ n-i ; i = 1,2...,L-1} along the surviving path and adding, as a last symbol, a phase difference Δ n at the time nT to the sequence so as to form a candidate sequence {Δ n-i ; i=0,1,2,...,L-1}, where L is a predetermined integer and L≥Q; and b-1-2) adding a detected phase Ψ n-q at a time (n-q)T to the sum of the phase differences of a partial sequence {Δ n-i ; i=0,1,...,q-1} of the candidate sequence so as to obtain an estimated value of the phase Ψ n and calculating the difference between the estimated value and the phase Ψ n so as to obtain a phase error µ n (q); said step b-2) includes the step of: adding the v-th power values of the absolute values of the phase error µ n (q) from q= 1 to L so as to obtain the M branch metrics that represent the likelihood of M branches from the M Q-1 states S n-1 at the time (n-1)T to a state S n at the time nT λ n = µ n (1) v + µ n (2) v + ... + µ n (L) v ; said step b-3) includes the step of: adding the branch metric λ(s n-1 →s n ) to a path metric Λ(S n-1 ) of the state S n-1 at the time (n-1)T that is read from the metric memory so as to obtain the path metric Λ(S n |S n-1 ) of the candidate sequences that pass through the state S n-1 , and comparing all of the M path metrics Λ(S n |S n-1 ) so as to obtain a state S n-1 ' with the minimum value; said step b-4) includes the steps of: b-4-1) defining the state S n-1 ' as a state at the time (n-1)T of a survival path leading to the state S n at the time nT, storing the state S n-1 ' in the path memory, defining the path metric Λ(S n |S n-1 ') as a path metric Λ(S n ) of the state S n at the time nT, and storing the path metric Λ(S n ) in the path metric memory; b-4-2) repeating the calculations at the above step for all the M Q-1 states at the time nT so as to obtain M Q-1 path metrics and comparing the M Q-1 path metrics so as to obtain a state S n ' with the minimum value; and b-4-3) tracing back the path memory for a predetermined interval DT starting from the state S n ', defining a phase difference Λ n-D that is one of Q-1 phase differences that constructs the state S n-D as a decoded symbol, and outputting the decoded symbol. The differential detection method as set forth in claim 3, wherein L = Q. The differential detection method as set forth in claim 3, wherein Q = 2. The differential detection method as set forth in claim 3, wherein L > Q. The differential detection method of claim 1, wherein the method uses a path memory and a path metric memory, the path memory being adapted for storing both, M states that represent modulated phase differences at each time and phase difference states at the just preceding time of maximum likelihood paths each leading to one of the M states at each time, the path metric memory being adapted for storing path metrics that represent the likelihood of sequences leading to each state, wherein: said step a) includes steps of: detecting a phase Ψ n of a received signal by quadrature differential detection and sampling a received signal at a time nT in a predetermined transmitted symbol interval T so as to obtain a received signal sample z n ; said step b-1) includes the steps of: b-1-1) tracing back the path memory starting from one state Δ n-1 of the M states at a time (n-1)T, obtaining a sequence {Δ n-i ; i=1,2,...,L-1} along a surviving path to the state Δ n-1 , adding a state Δ n and forming a candidate sequence {Δ n-i ; i=0,1,2,...,L-1}; b-1-2) rotating the phase of a received signal sample z n-q for the sum of the partial sequence {Δ n-i ; i=0,1,2,...,L-1} of the candidate sequence from q = 1 to L and adding L obtained values so as to obtain an estimated value z n ' of the received signal sample z n ; said step b-3) includes the steps of: b-3-1) calculating a real value of the inner product of the received signal sample z n and the estimated value z n ' and defining the real value as a branch metric λ(Δ n-1 →Δ n ) that represents the likelihood of transition from the state Δ n-1 at the time (n-1)T to the state Δ n at the time nT; b-3-2) adding the branch metric λ(Δ n-1 →Δ n ) to a path metric Λ(Δ n-1 ) of the state at the time (n-1)T so as to obtain a path metric Λ(Δ n |Δ n-1 ) of the candidate sequence that passes through the state Δ n-1 ; b-3-3) repeating the above calculations for all the M states Δ n-1 so as to obtain path metrics of the M candidate sequences and comparing the M candidate sequences so as to obtain a state Δ n-1 ' with the minimum value; b-3-4) deciding the state Δ n-1 ' as a state at the time (n-1)T of a surviving path to the state Δ n at the time nT, storing the state Δ n-1 ' in the path memory, defining the path metric Λ(Δ n |Δ n-1 ') as a path metric Λ(Δ n ) of the state Δ n at the time nT, and storing the path metric Λ(Δ n ) in the path metric memory; b-3-5) repeating the calculations at the steps b-1-1) to b-3-4) for all the M states at the time nT so as to obtain M path metrics and comparing the M path metrics so as to obtain a state Δ n ' with the minimum value; and said step b-4) includes the step of: tracing back the path memory for a predetermined interval DT starting from the state Δ n ' and outputting an obtained state Δ n-D as a decoded symbol. A differential phase detection method of an M-phase DPSK modulated signal, comprising the steps of: a) detecting a phase Ψ n of a received signal in a predetermined transmitted symbol interval T relative to a local signal; b) obtaining the difference Ψ n -Ψ n-q between the detected phase Ψ n and a phase Ψ n-q of up to L symbols before, where q = 1,2,..., L; c) obtaining the sum δ n-1 (q) = ΣΔ  n-i , where Σ is the sum of the phase differences from i = 1 to q-1 decided for each of up to (q-1) symbols before, and obtaining the v-th power value of the absolute value of the difference µ n (q) between the detected phase difference Ψ n -Ψ n-q and the sum of the added value δ n-q (q) and the phase difference candidate Δ n ' as a metric of the q-symbol differential phase detection; d) adding metrics of the L phase differences so as to obtain a branch metric λ n = |µ n (1)| v + ... + |µ n (L)| v for the phase difference candidate Δ n '; and e) outputting the phase difference candidate with the minimum branch metric as a decided phase difference Δ  n . A differential phase detector for successively estimating a sequence of phase differences at symbol intervals of a phase signal obtained by phase-detecting an M-phase DPSK modulated signal, comprising: local signal generating means (13) for generating a local signal with a frequency nearly the same as that of a transmitted signal modulated with M-phase DPSK at symbol intervals T; phase detecting means (12) for receiving the transmitted signal and detecting the phase of the received signal relative to the phase of the local signal; sampling means (14) for sampling the phase of the received signal every symbol interval T and outputting a sampled phase Ψ n ; delay means (15 1 -15 N ) having N delay stages connected in series and each causing delay for the symbol interval T, where N is an integer that is 2 or greater, said delay means being adapted for receiving the sampled phase from said sampling means and outputting sampled phases Ψ n-1 , Ψ n-2 ,...,Ψ n-N of 1 to N symbols before; and metric calculating means (16) for calculating metrics of sequences of phase differences; characterized in that said metric calculating means (16) is adapted for receiving the present sampled phase from said sampling means (14) and N past sampled phases from the delay stages of said delay means (15 1 -15 N ), adding the sum of a partial sequence {Δ i ; i=n, n-1,...,n+1-q} of q symbols before in an N-symbol phase difference sequence candidate {Δ n ; n=1,...,N} to a detected phase Ψ n-q of q symbols before so as to obtain an estimated value Ψ n ' of the phase Ψ n , defining the v-th power value of the absolute value of a difference µ n (q) between the estimated value Ψ n ' and the phase Ψ n as a metric of a q-symbol phase detection, where v is any real number that is 1 or greater, summing up the metrics for said estimated phase values to obtain a branch metric λ n for a path of said phase difference sequence candidate, summing up the branch metrics along said path to obtain a path metric A, and outputting an N-symbol phase difference sequence with the minimum path metric as a decoded sequence. The differential phase detector of claim 9, wherein said metric calculating means (16) calculates the branch metric λ n by λ n = µ n (1) v + µ n (2) v + ... + µ n (n) v , and sums up the branch metric from n = 1 to N so as to obtain a path metric Λ = λ 1 + λ 2 + ... + λ N of the phase difference sequence candidate {Δ i ; i=1,2,...,N}. The differential phase detector of claim 9, wherein said metric calculating means (16) includes a path memory (16B) and a path metric memory (16A), the path memory being adapted for successively storing the states of surviving paths at the just preceding time each leading to one of M Q-1 states defined by a sequence of past Q-1 phase difference, Q being a predetermined integer which is 2 or greater, the path metric memory being adapted for storing path metrics that represent the likelihood of surviving sequences each leading to one of M Q-1 states, said metric calculating means (16) being adapted for receiving (L-1) sampled phases Ψ n-1 , Ψ n-2 ,..., Ψ n-L at past times (n-1)T, (n-2)T,..., (n-L+1)T from the delay portions of said delay means and decoding the sampled phases based on the Viterbi algorithm with the M Q-1 states, wherein said metric calculating means (16) is adapted for adding the sum of a partial sequence {Δ i ; i=n, n-1,...,n+1-q} of q symbols of an L-symbol phase difference sequence candidate {Δ n ; n = 1,2,...,L} to a detected phase Ψ n-q of q symbols before so as to obtain an estimated value Ψ n ' of the phase Ψ n- , defining the v-th power value of the absolute value of a difference µ n (q) between the estimated value Ψ n ' and the phase Ψ n as a metric of q-symbol differential phase detection, where v is a real number that is 1 or greater, calculating a metric λ n (s n-1 →S n ) of M branches that leave M Q-1 states S n-1 = (Δ n-(Q-1) ,...,Δ n-2 , Δ n-1 ) at the time (n-1)T to each state S n = (Δ n-(Q-2) ,...,Δ n-1 , Δ n ) at the time nT according to the following equation λ n = µ n (1) v + µ n (2) v + ... + µ n (L) v , calculating path metrics of the M branches to the states S n according to the following equation Λ n (S n |S n-1 ) = Λ n-1 (S n-1 ) + λ n (S n-1 →S n ) defining a branch with the minimum path metric as a surviving path to the state S n ' storing the surviving path in the path memory, defining the value of the minimum path metric as a path metric Λ n (S n ) of the state S n , storing the path metric Λ n (S n ) in the metric memory, selecting a path leading to a state with the minimum path metric Λ n (S n ) of all the M Q-1 states S n at the time nT, tracing back states stored in the path memory along the selected path for a DT interval, and outputting a phase difference Δ n-D of states (Δ n-(D+Q-2) ,...,Δ n-(D+1) , Δ n-D ) as a decoded result Δ n-D '. A differential phase detector, comprising: local signal generating means (13) for generating a local signal with the same frequency as that of a received signal M-phase DPSK modulated at symbol intervals T; and phase detecting means (12, 14) for detecting the phase of the received signal relative to the local signal and outputting the detected phase Ψ n at the symbol intervals T; characterized by phase difference detecting means (15) for obtaining a phase difference Ψ n -Ψ n-q between the detected phase Ψ n and a phase Ψ n-q of up to L symbols before, where L is an integer that is 2 or greater; cumulating means (18) for cumulating decided phase differences δ n-1 (q) = ΣΔ  n-i where Σ is the sum of the phase differences from i=1 to q-1; and metric calculating means (16) for calculating the v-th power value of the absolute value of the difference between the obtained phase difference Ψ n -Ψ n-q and the sum of the added value δ n-1 (q) and a phase difference candidate Δ' as a metric λ q of a q-symbol differential phase detection, adding L metrics of the differential phase detection so as to obtain a branch metric λ n = |µ n (1)| v +...+|µ n (L)| v and outputting a phase difference candidate with the minimum branch metric as a decided phase difference Δ  n . The differential phase detector as set forth in claim 12, wherein said phase difference detecting means (15) includes delay means (15 1 -15 N ) having L delay stages and L subtracters (15S 1 -15S L ), the L delay stages being connected in series and each causing delay for a symbol interval T, the L subtracters being adapted for successively receiving detected phases from said phase detecting means and outputting past detected phases Ψ n-1 , Ψ n-2 ,..., Ψ n-L of up to 1 to L symbols before, the subtracters being adapted for subtracting the output phases of the delay stages from the detected phase Ψ n received from said phase detecting means and outputting the phase difference Ψ n -Ψ n-q . The differential phase detector as set forth in claim 12, wherein said cumulating means (18) has delay means (18 1 -18 L-1 ) and (L-2) adders (18A 1 -18A L-2 ), the delay means having (L-1) delay stages connected in series and each causing delay for a symbol interval T, the adders being adapted for receiving successively decided phase differences from said metric calculating means, receiving past decided phase differences Δ  n-1 , Δ  n-2 , Δ  n-L of 1 to L symbols before from the delay stages, cumulating the output phases of the delay stages, and generating a sum δ n-1 (q) = ΣΔ  n-1 from i = 1 to q-1 and from q =2 to L.

Inventors: ADACHI FUMIYUKI, DOHI TOMOHIRO, SAWAHASHI MAMORU, ADACHI, FUMIYUKI, DOHI, TOMOHIRO, SAWAHASHI, MAMORU
Applicants: NIPPON TELEGRAPH & TELEPHONE, NTT MOBILE COMMUNICATIONS NETWORK INC.
Application year: 1994
Main classification: H04L27
IPC codes: H03M13, H04L25, H04L27