Voltage multiplier for high output current with stabilized output voltage

Abstract: A voltage multiplier for relatively high output current has its design output voltage stabilized and rendered independent of process spread, temperature, supply voltage and output current level, by a stabilization loop driving the switch that cyclically connects to ground a charge transfer capacitance of the functional voltage multiplier circuit. The feedback loop comprises an integrating stage, stabilized by creating a low-frequency zero in the transfer function for compensating one of two low-frequency poles of the transfer function of the whole circuit.

Description: The present invention concerns a charge pumping circuit (e.g. voltage multiplier or voltage booster), which is provided with an innovative arrangement for stabilizing its output voltage. Often in electronic systems there is a need of generating a DC voltage higher than the supply voltage. Such a voltage boosting function is commonly effected with a so-called voltage multiplier or voltage booster. However, the circuit implementations of a voltage booster may be differ considerably depending on the level of the current that the circuit should be able to deliver. For relatively low output current levels (in the order of 10 µA), as in the case of write operations of EPROM memories, an often used solution is the circuit shown in Fig. 1a (employing bipolar junction transistors BJT) or in Fig. 1b (employing MOS transistors), as described in Witters et al ., "Analysis and Modelling of On-Chip High-Voltage Generator Circuits for Use in EEPROM Circuits", 24 IEEE JOURNAL OF SOLID-STATE CIRCUITS 1372 (October 1989), which is hereby incorporated by reference. Conversely, when the voltage multiplier must deliver output currents of a remarkable level (in the order of ten mA), as in the case of analog integrated circuits wherein the supply voltage must be doubled or tripled for attaining certain performances, a typical solution requires the employment of at least two large capacitances (normally external to the integration circuit). A first capacitor C 1 is used for loading and transfering electric charge, while the second capacitor C 2 has the function of storing the periodically transferred charge, as schematically depicted in Fig. 2. The four switches (SW1, SW2, SW3, SW4) are driven by first and second phase control signals ( 1 and  2 ). Basically, during a first phase ( 1 ) of the control clock, the capacitance C 1 is connected between V CC and ground (GND) and therefore a voltage equal to V CC develops on the "upper" terminal as referred to the "lower" terminal of the capacitor that is connected to ground. During a second phase ( 2 ) of the control clock, the lower terminal is connected to V CC while the upper terminal is connected to the output node (on C 2 ), onto which a voltage V sur develops. If the output current is null, then it is easy to see that, after several clock periods, the output voltage V sur would become equal to 2V CC and perfectly stable. The output capacitance C 2 stores the output voltage V sur . When a load connected to the output node (V sur ) draws current, the resulting value of the output voltage V sur will be determined by the balance between the current drawn by the load and the quantity of charge per unit time delivered through the transfer capacitance C 1 , while the output capacitance C 2 limits the amplitude of the "ripple" of the output voltage V su r. In Callias et al ., "A Set of Four IC's in CMOS Technology for a Programmable Hearing Aid", 24 IEEE JOURNAL OF SOLID-STATE CIRCUITS 301 (April 1989), which is hereby incorporated by reference, it is explained that the value of the output voltage V sur in case of current absorption by a load, may be easily calculated by analyzing the equivalent circuit shown in Fig. 3. 1 By following such a derivation, the value of the output voltage in the circuit of Figure 2 may be seen to be given by the following relationship: V sur = 2 V CC - ( R C1 +R τ ) I L where R C1 and R τ are respectively given by the following equations: R C1 = 1 ƒ s C 1 R τ = 1 ƒ S C A [ e - t / t 1 1- e - t / t 1 + e - t / t 2 1- e - t / t 2 where C A is the capacitance of C 1 and C 2 in series, and t 1 and t 2 are RC time constants: C A = C 1 C 2 C 1+ C 2 t 1 = ( r ON1 + r ON2 ) C 1 t 2 = ( r ON3 + r ON4 ) C A In the equations above, f s is the switching frequency of the control clock, and the r ON values represent the respective internal resistance of the switches SW1, SW2, SW3 and SW4, that are employed in the circuit. As may be easily observed, the effective value of the output voltage V sur depends strongly not only on the current I L absorbed by the load and the value of the supply voltage V CC , but also on the internal resistances r ON of the transistors that implement the switches SW1, SW2, SW3 and SW4, that drive the circuit. The on-resistance of a transistor is an intrinsic parameter which is notoriously dependent from the fabrication process, temperature, supply voltage, etc. Many high precision analog circuits cannot tolerate an excessive spread of the actual output voltage (V sur ) from the design value. Thus, a main object of the present invention is to provide a simple low-power voltage multiplier or voltage booster, designed for a relatively high output current, the output voltage of which is stable and substantially independent from process spread, temperature, supply voltage and, within certain limits, also from the output current that is actually delivered to a load. This object and other advantages are achieved by the output voltage stabilizing circuit of the present invention. The circuit of the invention employs an integrating stage, capable of generating a DC error signal, representative of the difference between a reference voltage and the output voltage of the voltage multiplier. The error signal is employed for driving (during a conduction phase of the operating cycle of the multiplier) a transistor that acts as a switch for connecting to ground at least a charge transfer capacitance (C 1 ) of the voltage multiplier. The generated error signal effectively controls the internal resistance r ON , so as to keep constant the output voltage V sur of the voltage multiplier. Such a driving error signal is fed to a control terminal of the transistor (switch) that connects to ground the charge transfer capacitance of at least a stage of the voltage multiplier, under control of a pair of timing signals, which control a transfer gate of the signal and an auxiliary switch for discharging the control node of the transistor (switch) driven by the error signal. In the preferred CMOS implementation, PMOS transistors are used to connect the upper terminal of the charge transfer capacitor alternately to the positive supply voltage Vcc and the pumped voltage Vsur, and overdriven clock voltages are used to assure reliable turn-off of these transistors. Some attempts have been made to provide regulation of a pumped voltage. European application EP-A-0540948 of Zavaleta (Motorola) uses a feedback loop to act on the r ON of a MOS transistor for regulating the output of the charge pump at the desired value. However, the presently preferred embodiment of the present application provides advantages over the Motorola circuit in the manner of implementing the feedback loop. As may be observed from Fig. 4 of the Motorola application, the circuit that implements a proportional + integral function is very complex. It uses three amplifiers (120, 130 and 142), three capacitances (137, 135 and 139), and several switches. By contrast, the innovative circuit disclosed in the present application provides a very simple structure composed of a single amplifier, a capacitance and a resistance for implementing the same function. Moreover, since the Motorola circuit separates the proportional function from the integral function pair, they are obliged to drive two PMOS transistors (150 and 151) in order to vary the r ON ; but in the present invention the two functions are implemented together, and in the presently preferred embodiment driving a single n-channel MOS suffices. Last but not least, given that a main objective in battery powered systems is to limit power consumption as much as possible for prolonging the life of the battery, the innovative circuit of the presently preferred embodiment consumes much less current (power) than the Motorola's circuit. The disclosed circuit employs a single amplifier directly powered by the battery, while Motorola utilizes three amplifiers, of which at least two (120 and 142) are powered at high (boosted) voltage, that is they consume at least five times the current that is consumed in the innovative circuit of the presently preferred embodiment for implementing the feedback loop. In this connection, it should also be noted that the Motorola circuit which generates PTUBBIAS, which biases the n-wells of the PMOS 150, 151, 152 and 154 at the highest potential of VBAT and VDD in order to avoid direct biasing, does not satisfy at all the function that is required at power-up. In fact, by supposing that at power-down VDD = 0V and that therefore PTUBBIAS = VBAT, when leaving the power-down condition, the capacitance 80 will have charged to VBAT during 1 and therefore, during 2, will tend to strongly pull-up the node 72, thus forward biasing the diode from p + (node 72) to n-well (common to 150, 151 and 152). This could give rise to a sudden current absorption in the order of 10-60 mA from VBAT, unduly degrading the battery. The improved circuit embodiments of Figures 6 and 7 provide an additional protection against a current surge when power is first applied, by including small resistors in series with the wells of the devices which are interposed between Vcc and the pumped node V sur . Preferably these resistors are formed merely by lateral extensions of the n-well diffusion. The invention is defined in the annexed claims 1 and 7, in terms of a novel voltage multiplier circuit and in terms of a novel method to generate a multiplied voltage, respectively. Preferred embodiments are recited in dependent claims 2-6 and 8. The disclosed inventions will be described with reference to the accompanying drawings, which show important sample embodiments of the invention and which are incorporated in the specification hereof by reference, wherein: The different aspects and advantages of the circuit of the invention will be more easily understood through the following description of an important embodiment and by referring to the annexed drawings, wherein: Figure 1a is a circuit diagram of a voltage multiplier for low output current, made with bipolar junction components, as described above; Figure 1b is a circuit diagram of a voltage multiplier for low output current, made with MOS components, as already described above; Figure 2 is a functional diagram of a voltage multiplier suitable to deliver a relatively high output current; Figure 3 is an equivalent circuit of the functional diagram of Fig. 2; Figure 4 is a functional diagram of a voltage multiplier provided with means for stabilizing the output voltage, according to the present invention; Figure 5 is a timing diagram of the various signals of the circuit of the invention; Figure 6 shows an alternative embodiment which has now been discovered to be advantageous; Figure 7 shows a further alternative embodiment in a P-well CMOS process; and Figure 8 shows voltage waveforms for the embodiment of Figure 7, and signal levels during power-down. The numerous innovative teachings of the present application will be described with particular reference to the presently preferred embodiment (by way of example, and not of limitation), in which: By referring to the circuit diagram of Fig. 4, the feedback loop for stabilizing the output voltage of the circuit according to the presently preferred embodiment of the present invention, may be structured as follows. A voltage divider, which in the example shown, is represented by the two resistances R1 and R2 connected in series between the output node V sur of the voltage multiplier circuit and a common ground node of the circuit, provides on an intermediate node thereof, an attenuated replica of the output voltage V sur of the voltage multiplier. This replica signal is fed to an inverting input of an operational amplifier OP-AMP, which, being powered at V CC , could not otherwise tolerate an input voltage higher than V CC . An integrating stage is constituted by the operational amplifier OP-AMP, provided with a feedback loop that essentially comprises an integrating capacitance C I . The integrating stage practically compares the replica signal V x , provided by the voltage divider R1-R2, with a reference voltage V RIF and outputs a DC error signal. A switch or "transfer gate", which, as in the example shown, may be composed of a MOS transistor pair MA-MB (where MA is PMOS and MB NMOS), is controlled by a first clock phase  1 and by its inverted replica  1 \. The transfer gate MA-MB, during a first phase  1 of the control clock, transfers the error signal generated by the integrating stage to the control gate of the NMOS trnasistor M1, which constitutes the switch SW1 of the functional circuit of the voltage multiplier (Fig. 2). A third MOS transistor MC is activated by the clock's phase  2 and during this phase of the clock's cycle, connects to ground the gate of M1, thus ensuring its cut-off. By virtue of the high DC gain of the stabilizing feedback loop, the feedback forces the condition: V x = V RIF and therefore: V sur = V RIF R 1 + R 2 R 1 It should be immediately noted from the above equation that the output voltage V sur is independent from process spreads, temperature, V CC and output current delivered to a load. A resistance R z , connected in series with the integrating capacitance C I of the feedback line of the operational amplifier OP-AMP, has an important function for ensuring a great stability of the circuit. In fact, the circuit presents two low-frequency "poles" (the DC pole of the integrating stage and a second low-frequency pole intrinsic to the equivalent circuit of the voltage multiplier circuit, as depicted in Fig. 3) that would tend to restrict the stability margin. The introduction of a resistance R z , in series with C I in the feedback line, determines the creation of a low-frequency "zero" that compensates a low-frequency "pole", thus enhancing stability. The positive effects of the resistance R z may be observed also during the power-on transient of the circuit, during which the system operates in a nonlinear manner. The presence of a series resistance R z reduces the "settling" time (the time that the circuit employs from the power-on instant to reach a steady state condition, that is for developing the design voltage V sur on its output node) by a factor that may be comprised between 10 and 1000, depending upon operating conditions and structural parameters of the circuit (process, temperature, output current, V CC ). A timing diagram showing the waveform of the control signals employed in the circuit of Fig. 4, according to an embodiment of the invention, are shown in Fig. 5. As may be seen, the phase  1 ( 1 and  1 \) is generally coincident with the phase  1sur , while the phase  2 ( 2 ,  2 \) is generally coincident with the phase  2sur . However, note that this is NOT true of  2sur \ in the power-down condition: in the power-down condition both  1sur \ and  2sur \ are low, and therefore both M2 and M4 will turn on as Vcc rises. The phases  1sur \ and  2sur \ may be generated, e.g. , by a timing circuit powered at the output voltage V sur of the voltage multiplier. In other words, the driving phases  1sur \ and  2sur \ are voltage-boosted phases. The other switches M3 (PMOS), MC, MA and MB may be controlled by control phases that are generated by a timing circuit similar to the one employed for generating the boosted phases  1sur \ and  2sur \, but powered at the supply voltage V CC . This is not strictly essential for the operation of the circuit, and, in practice, also the switches M3, MC, MA and MB may be controlled by employing the same voltage boosted phases  1sur \- 1sur and  2sur \- 2sur , as will be evident to a person skilled in the art. In the sample implementation of Figure 4, note that transistors M2, M3, and M4 are all PMOS. This helps to avoid incurring V T drops from the supply voltage V CC . However, of course, one of ordinary skill in the art can implement the disclosed invention in various other circuit arrangements, and in technologies other than CMOS. Figure 6 shows an alternative embodiment which has been discovered to be advantageous. The circuit of Figure 4, wherein the n-well of two PMOS devices (M1 and M4) have been joined to the respective sources, may still exhibit a current spike at power-up. Figure 6 shows a further improvement of this circuit, including an expedient for reducing drastically the current spike at power-up. In a power-down condition M1, M2, M4 are ON and M3 is OFF, as shown at the left side of Figure 5. (The clocks are conditioned on the power-down signal PD, which is used to reduce to zero the current consumption of the IC and of the voltage multiplier when the IC itself is not used). Moreover, during power-down, the added MOS MPD (gated by complemented power-down signal PD\) is OFF (and therefore prevents current consumption by the voltage divider R1+R2), while the amplifier turns off, forcing its output to Vcc voltage (thus confirming ON the transistor M1). In this embodiment, two resistors Rb2 and Rb4 have been inserted in series with the n-wells of M2 and M4, respectively, in order to limit the current spikes at power-up. From simulation, it has been found that a resistance of a few KΩ, which may be realized with the n-well cell itself, will be sufficient to reduce the injection of current by a factor of 100, that is to a few tenths of a milliAmp, which should be readily tolerated by the battery. Note that the resistors Rb2 and Rb4 do not pass any significant current during normal operation: the importance of these resistors is primarily at start-up. Figure 7 shows a further alternative embodiment in which charge pumping is used to achieve a negative voltage V sur ' from a positive supply, in a P-well CMOS process. In this embodiment the polarities and connections of many of the transistors have been reversed, and accordingly the reference numerals have been primed. Thus PMOS M1' in Fig. 7 is analogous to NMOS M1 in Fig. 6, NMOS M2' in Fig. 7 is analogous to PMOS M2 in Fig. 6, NMOS M3' in Fig. 7 is analogous to PMOS M3 in Fig. 6, NMOS M4' in Fig. 7 is analogous to PMOS M4 in Fig. 6, PMOS MC' in Fig. 7 is analogous to NMOS MC in Fig. 6, PMOS MB' of Fig. 7 is analogous to NMOS MB of Fig. 6 and NMOS MA' of Fig. 7 is analogous to PMOS MA of Fig. 6. , A similar polarity-reversal is applied to the clock signals: note that  1 in Figure 6 is replaced by its complement  1 \ in Figure 7, and  2 is replaced by its complement  2 \. Moreover, the overdriven positive clocks  1sur and  2sur are replaced by negatively overdriven (sub-ground) clocks  1sur ' and  2sur '. As in Figure 6, the resistors Rb2 and Rb4 limit the current surge caused by shifting the potential of the wells when power is first applied. The timing diagram of Figure 8 shows the clock voltage levels in the circuit of Figure 7, in power-down and in operating conditions. Note that in the power-down condition transistors M2' and M4' are both ON. Of course, these alternatives are merely a few of the many possible modifications and variations which may be made to the innovative ideas as disclosed in claim 1.

Claims: A voltage multiplier circuit, comprising: at least a first charge transfer capacitance (C1) and a second output storing capacitance (C2); a first switch (M1), for connecting to ground a first terminal of said first capacitance (C1); said first terminal of said first capacitance being connected to ground only through said first switch; a second switch (M2) for connecting a second terminal of said first capacitance (C1) to a supply node (Vcc); a third switch (M3) for connecting said first terminal of said first capacitance (C1) to said supply node; a fourth switch (M4) for connecting the second terminal of said first capacitance (C1) to a first terminal of said second capacitance (C2) which provides an output node of the voltage multiplier; said second switch (M2) being controlled by a first clock phase (Φ1 sur ) and said third (M3) and fourth (M4) switches being controlled by a second clock phase (Φ2); and further comprising a voltage divider (R1, R2) functionally connected between said output node and ground; a differential amplifier (OP AMP) having a noninverting input (+) to which a reference voltage (Vrif) is fed, an inverting input (-) connected to an intermediate node of said voltage divider (R1, R2) and to the output of the amplifier (OP AMP) through a series comprising an integrating capacitance (CI) and a resistance (Rz); a first transfer gate (MA, MB), controlled by a first timing signal (Φ1) and by a complementary signal (Φ1), coincident with said first clock phase (Φ1 sur ), an input of said transfer gate being connected to said output of the amplifier and an output of said transfer gate being connected to a control terminal of said first switch (M1); a fifth switch (MC), controlled by a second timing signal (Φ2) coincident with said second clock phase (Φ2) being functionally connected between said control terminal of said first switch (M1) and ground. The circuit of claim 1, further comprising one or more resistors (R2, R4) interconnected with said second (M2) and fourth (M4) switches to limit transient current between said first supply node (Vcc) and said output capacitor (C2), when power is first applied to said first supply node. The circuit of claim 1, wherein said first switch (M1) consists of an N-channel insulated-gate filed-effect transistor. The circuit of claim 1, wherein said second (M2), third (M3), and fourth (M4) switches each consist of a P-channel insulated-gate field-effect transistor. The circuit of claim 1, wherein said second (M2) and fourth (M4) switches each consist of a P-channel insulated-gate filed-effect transistor, and are connected to be driven by a clock voltage (Φ1 sur , Φ2 sur ) which is at least approximately equal to the multiplied voltage. The circuit of claim 1, wherein said transfer gate (MA, MB) consists of a P-channel insulated-gate field-effect transistor connected in parallel with an N-channel insulated-gate filed-effect transistor. A method for generating from first and second supply voltages (Vcc, GND), using at least one charge transfer capacitor (C1), a pumped voltage (Vsur), on an output storing capacitor (C2), which is not intermediate between said first and second supply voltages, comprising the steps of: a) on a first clock phase (Φ1), connecting a first terminal of said charge transfer capacitor (C1) to said first supply voltage (Vcc), and connecting a second terminal of said charge transfer capacitor to said second supply voltage (GND), using at least one variable impedance element (M1) which is regulated in accordance with variations of said pumped voltage (Vsur) from a target voltage, b) on a second clock phase (Φ2) which does not overlap with said first clock phase (Φ1), connecting (M3) said second terminal of said charge transfer capacitor (C1) to said first supply voltage (Vcc), and connecting (M4) said first terminal of said charge transfer capacitor to provide said pumped voltage (Vsur) on said output storing capacitor (C2), said steps a) and b) being repeatedly alternately, characterized by comprising the additional step of dividing down said pumped voltage, using a voltage divider (R1, R2) and disabling it during a power-down condition, to provide a scaled voltage (Vx) comparing the scaled voltage against a reference voltage (Vrif) and controlling said variable impedance element (M1) accordingly. The method of claim 7, further comprising the step of routing at least some of the transient current between said first power supply and said output storing capacitor through one or more series resistors, when power is first applied.

Inventors: CONFALONIERI PIERANGELO, CRIPPA CARLO, NICOLLINI GERMANO, CONFALONIERI, PIERANGELO, CRIPPA, CARLO, NICOLLINI, GERMANO
Applicants: ST MICROELECTRONICS SRL, STMICROELECTRONICS S.R.L.
Application year: 1994
Main classification: H02M3
IPC codes: H02M3