Brushless DC motor

Abstract: A brushless DC motor comprises a frequency generator (1) which generates plural-phase frequency signals (m₁,m₂) proportional to a rotational speed of a rotor (20), a direction detecting circuit (3) which detects a rotational direction of the rotor from the plural-phase frequency signals to output a direction signal (d), a counting circuit (4) which up-counts or down-counts pulses of a frequency signal in response to the direction signal, an initial position detecting circuit (8) which provides an initial value to the counting circuit, a phase detecting circuit (10) which detects the phase of a current or voltage to be conducted to the plural-phase stator windings (11,12,13) to output a phase signal, a phase adjusting circuit (5) which adjusts a count value of the counting circuit in response to the phase signal, a waveform generating circuit (6) which generates plural-phase waveform signals in response to an output of the phase adjusting circuit, and a power supply circuit (7) which supplies voltages or currents to the stator windings respectively in accordance with the plural-phase waveform signals. In addition, the phases of the voltages induced respectively in the stator windings and the phases of the driving currents are corrected so as to be coincided respectively with each other when the motor is being rotated, so that the motor can be always driven at a highest efficiency.

Description: BACKGROUND OF THE INVENTION 1. Field of the Invention This invention relates to a brushless DC motor which does not have a position sensor for detecting a rotational position of a permanent magnet rotor. 2. Description of the Prior Art Brushless DC motors are longer in service life and less in noise generation for the reason that they do not need to have a mechanical contact as would be used in conventional DC motors having a brush. As a result, they have been widely used recently in industrial or audio and video equipment requiring higher reliability (see for example EP-A-0313046). In order to perform the switching operation of a conducting phase of the stator windings of a motor, most of conventional brushless DC motors use a rotor position sensor (such as, for example, the Hall sensor) instead of a brush. However, the rotor position sensor is not so cheap, it requires sophisticated positional adjustment for setting and increased number of wirings, so that the cost of the brushless DC motors is large as compared with DC motors having a brush. Besides, some structural limitations will be frequently imposed thereupon for the reason that a rotor position sensor has to be set inside the motor itself. A recent trend is that accompanied with the compactization of industrial or audio and video equipment, motors to be used are made smail in size and thickness, which means that the sectional space where a motor position sensor such as the Hall sensor is to be provided becomes extremely small. Thus, several types of brushless DC motor having no position sensor such as, for example, the Hall sensor have been proposed previously. Out of which, such a brushless DC motor is known that uses an output pulse of a frequency generator fixed to a motor. This motor counts an output pulse of the frequency generator which generates or pulse in response to the rotation of the rotor by a counter circuit and conducts a driving current having a current pattern preset in response to the count value thus obtained to the three-phase stator windings of the rotor in a successive manner thereby rotating the permanent magnet rotor (see, for example, the Japanese Patent Application Laid-Open No. 63-262088). With the structure as explained above, the initial position of the rotor when starting cannot be found. As a result, tne conventional brushless DC motor as shown above has a reset signal generating circuit particularly to reset the counter using a reset signal when starting and supply a specific reset signal simultaneously to the stator windings of the rotor, whereby the rotor and the stator windings are made in a predetermined positional relation to each other in advance. However, if the specific reset current is supplied to the stator windings in order to determine the initial position, the rotor is started to rotate and the position of the rotor becomes vibrative around the predetermined position, so that it is impossible to stabilize to the predetermined position in a short period of time. As a result, it is impossible to shift the made in a short period of time from the reset mode in which the specific reset current is supplied to the stator windings to stabilize the rotor to the predetermined position when starting to the normal position detecting mode in which an output pulse of the frequency generator is counted in response to the rotation of the rotor. Thus, there arises such a problem that the starting time becomes long. This means that a conventional brushless DC motor cannot be used in such an application that it is required to have rotation and stop frequently repeated and to be started in a short period of time. In addition, with the conventional brushless DC motor as explained above, the initial position of the rotor is detected when starting, so that even if the rotor and stator windings are made in a predetermined positional relation to each other by supplying the reset current to the stator windings, when the rotor is applied with a load, the positional relation between the rotor and stator windings will be largely varied depending on the magnitude of the load thus applied. As a result, it becomes impossible to stabilize the rotor at the predetermined position in the reset mode. Accordingly, with the conventional brushless DC motor as shown above, if the mode is shifted from the reset one to the normal position detection one for counting the output pulse of the frequency generator in response to the rotation of the rotor, the phase of a current supplied to the stator windings is largely deviated from the normal phase, resulting in being impossible to realize a highly efficient drive. As a result, a conventional brushless DC motor is disadvantageous in that it can be used only in such an application that the motor itself is unloaded when starting. In addition, with the brushless DC motor in which, as shown above, an output pulse of the frequency generator for generating a pulse in response to the rotation of the rotor is counted by the counter circuit and a driving current is conducted to the three-phase stator windings in a successive manner in response to the count value thus obtained thereby to rotate the permanent magnet rotor, the count value obtained by the counter circuit will become erroneous if the output of the frequency generator has a noise overlapped by any reason during continuous driving, and such count error thus generated leads to a reduction in efficiency of the motor and an increase in torque ripple. In addition, the accumulation of such count errors may cause, the motor to be stopped in the worst case, which means that the conventional brushless DC motor as shown above lacks reliability. SUMMARY OF THE INVENTION An object of this invention is to provide a brushless DC motor which is capable of detecting a positional relation of the rotor and the stator windings in a short period of time when starting the motor and which is capable of immediately switching the mode from a phase coinciding mode at the time of starting to a normal position detecting mode performed by counting output pulses in response to a rotational speed of the rotor. Another object of this invention is to provide a brushless DC motor which is capable of detecting a position of the rotor at an extremely high accuracy independently of the magnitude of a load even when the motor is already being subjected to an application of such load at the time of starting. Further another object of this invention is to provide a brushless DC motor which is capable of being driven always at a high efficiency by applying phase correction process so as to coincide the phase of each of the voltages induced respectively in the stator windings with the corresponding one of the driving currents conducted respectively to the stator windings when the motor is being rotated normally. In order to attain the above-mentioned objects, a brushless DC motor of this invention comprises a frequency generator for generating plural-phase frequency signals each having a frequency proportional to a rotational speed of a permanent magnet rotor, direction detecting means for detecting a rotational direction of the permanent magnet rotor from the plural-phase frequency signals to obtain a direction signal, counting means responsive to said direction signal for up-counting or down-counting pulses of at least one frequency signal of the plural-phase frequency signals, phase adjusting means for obtaining a command value from a count value of said counting means, waveform generating means for generating plural-phase waveform signals in response to said command value power supply means for supplying driving currents or driving voltages to the plural-phase stator windings in response to said plural-phase waveform signals. Phase detecting means outputs a phase signal in response to a phase of a voltage or current of at least one of said plural-phase stator windings. windings. Phase adjusting means adjusts a count value of said counting means by adding thereto or substracting therefrom a predetermined value in response to a rotational direction command and performs phase adjustment in response to said phase signal to obtain the command value. When starting the motor, by rotating the rotating magnetic field generated in the stator windings clockwise or counterclockwise, the initial phase of the permanent magnet rotor is coincided therewith, and when rotating the rotor normally, the phases of respective driving currents conducted to the stator windings and the phases of respective induce voltages induced in the stator windings are corrected so as to be always coincided with each other. With the structure as shown above, the brushless DC motor of this invention counts output pulses outputted from the frequency generator by the counting circuit. The position signal is formed based on the count value thus obtained and as a result, it does not need to use a position sensor that a conventional brushless DC motor requires to use. Consequently, it becomes possible to eliminate a sophisticated setting position adjustment of the sensor as well as to reduce the number of wirings, resulting in a vast reduction in cost. In addition, since it is unnecessary to provide a rotor position sensor into the motor, no structural limitation can be imposed thereupon, thus making possible to be small in size and thickness. Further in addition, the driving currents conducted respectively to the stator windings and the induce voltages induced respectively in the stator windings are subjected to correction so as to be always coincided in phase with each other, which means that the motor can be driven at a high efficiency. BRIEF DESCRIPTION OF THE DRAWINGS Fig. 1 is a block diagram of a brushless DC motor according to one embodiment of this invention, Fig. 2 is a circuit diagram of the direction detecting circuit shown in Fig. 1, Fig. 3 is a diagram showing a signal waveform at each member of the direction detecting circuit shown in Fig. 2. Fig. 4 is a circuit diagram of the phase detecting circuit shown in Fig. 1. Fig. 5 is a diagram showing a signal waveform at each member of the phase detecting circuit shown in Fig. 4. Fig. 6 is a diagram showing a signal waveform at each member of the circuit shown in Fig. 1 when rotating stationarily. Fig. 7 is a vector diagram showing a relation of a magnetic pole vector Φ of a permanent magnetic rotor, a magnetomotive force vector I generated by stator windings and an induce voltage E induced in the stator windings of the brushless DC motor shown in Fig. 1 when rotating stationarily. Fig. 8 is a block diagram showing an arrangement of a phase adjusting circuit, waveform generating circuit, initial position detecting circuit and selecting circuit making the brushless DC motor shown in Fig. 1. Fig. 9 is a flow diagram for explaining a normal mode process of the brushless DC motor shown in Fig. 1. Fig. 10 is a vector diagram showing a relation of a magnetic pole vector Φ of a permanent magnet rotor and a magnetomotive force vector I generated by stator windings of the brushless DC motor shown in Fig. 1 when phase coinciding operation. Fig. 11 is a flow diagram for explaining a phase coinciding operation of the brushless DC motor shown in Fig. 1. Fig. 12 is a diagram showing a signal waveform at each member of the brushless DC motor shown in Fig. 1 when the phase detecting circuit shown in Fig. 4 is used and the phase of a voltage induced in each of the stator windings is shifted from the phase of a driving current conducted to the corresponding one of the stator windings. Fig. 13 is a flow diagram for explaining a phase correcting operation of the brushless DC motor shown in Fig. 1. Fig. 14 is a circuit diagram of another embodiment of a phase detecting circuit shown in Fig. 4. Fig. 15 is a diagram showing a signal waveform of each member of the phase detecting circuit shown in Fig. 14. Fig. 16 is a diagram showing a signal waveform at each member of the brushless DC motor shown in Fig. 1 when the phase detecting circuit shown in Fig. 14 is used and the phase of a voltage induced in each of the stator windings is shifted from the phase of a driving current conducted to the corresponding one of the stator windings. Fig. 17 is a circuit diagram of another embodiment of a phase detecting circuit shown in Fig. 4. Fig. 18 is a diagram showing a signal waveform at each member of the circuit shown in Fig. 17. Fig. 19 is a circuit diagram of another embodiment of a power supply circuit shown in Fig. 1. Fig. 20 is a diagram showing a signal waveform at each member of the power supply circuit shown in Fig. 19. DESCRIPTION OF THE PREFERRED EMBODIMENTS Fig. 1 is a block diagram of a brushless DC motor according to one embodiment of this invention. In Fig. 1, 20 is a permanent magnet rotor, 11, 12 and 13 are three-phase stator windings and 1 is a frequency generator which generates two-phase frequency signals m1 and m2 different in phase from each other in proportional to the rotation of the permanent magnet rotor 20. The two-phase frequency signals m1 and m2 are sent to a waveform shaping circuit 2 to be converted respectively into rectangular wave signals s1 and s2 and then, inputted to a direction detecting circuit 3. The direction detecting circuit 3 outputs a direction signal d in response to the rotational direction normal or opposite of the permanent magnet rotor 20. In addition, 4 is a counting circuit which receives the rectangular wave signal s1 outputted from the waveform shaping circuit 2 and the direction signal d outputted from the direction detecting circuit 3 thereby to up-count or down-count the pulse-number of the rectangular wave signal s1 generated through the waveform shaping circuit 3 proportionally to the rotation of the permanent magnet rotor 20 in response to the rotational direction thereof, and 8 is an initial position detecting circuit which receives a count value c of the counter 4 to determine the initial position of the permanent magnet rotor by rotating the rotating magnetic field that the stator generates in the normal or opposite direction when the phases are coincided with each other and outputs an initial value q to the counting circuit 4. The initial position detecting circuit 8 also outputs an address command signal b to a selecting circuit 9 simultaneously when the phases are coincided with each other. A phase detecting circuit 10 compares the magnitudes of three-phase voltages applied respectively to the stator windings 11, 12 and 13 to output a phase signal g. In addition, 5 is a phase adjusting circuit which adds or subtracts a predetermined value to or from the count value c of the counting circuit 4 in response to a direction command r inputted to an input terminal 14, then, performs the phase correction in response to the phase signal g outputted from the phase detecting circuit 10 and outputs an address signal a to the selecting circuit 9. The selecting circuit 9 selects the address signal a or the address command b in response to a phase coinciding command t inputted to a terminal 15 thereby to output an address signal f to a waveform generating circuit 6. The waveform generating circuit 6 outputs three-phase position signals p1, p2 and p3 to a power supply circuit 7 in response to the address signal f outputted from the selecting circuit 9. The power supply circuit 7 amplifies the three-phase position signals p1, p2 and p3 thus inputted thereby to supply electric currents i1 , i2 and i3 which are proportional to the magnitudes of the position signals p1, p2 and p3 to the stator windings 11, 12 and 13, respectively. With the brushless DC motor structured as above, the operation will be explained in detail. First, the explanations will be made on the case that the permanent magnet rotor 20 is being rotated normally. Fig. 2 shows a circuit structure of a direction detecting circuit showing an example of the circuit 3 shown in Fig. 1, and Fig. 3 shows a signal waveform at each member of the circuit shown in Fig. 2. In Fig. 2, 21 is a D-type flip-flop circuit which receives the two-phase rectangular wave signals s1 and s2 outputted from the waveform shaping circuit 2. In the flip-flop circuit 21, the rectangular wave signal s1 is inputted to a data input terminal D and the rectangular wave signal s2 is inputted to a clock input terminal CK. Fig. 3a shows the waveforms of the rectangular wave signals s1 and s2 when the permanent magnet rotor 20 is being rotated normally, and Fig. 3b shows the waveforms of the rectangular wave signals s1 and s2 when it is being rotated opposite thereto. The D-type flip-flop circuit 21 holds the state of the data input terminal d at each rising edge of the signal inputted to the clock input terminal CK to output the state thereof from an output terminal Q and as a result, when the permanent magnet rotor 20 is being rotated normally, an output signal Q of the D-type flip-flop circuit 21 always becomes a high potential state (hereinafter called "H" state) as shown in Fig. 3a. In the other hand, when the permanent magnet rotor 20 is being rotated opposite, the output signal Q of the flip-flop circuit 21 always becomes a low potential state (hereinafter called "L" state) because the rectangular wave signal s1 is delayed in phase by 90° from the rectangular wave signal s2 as shown in Fig. 3b. As clear from the above explanations, a rotational direction of the permanent magnetic rotor 20 can be detected by the direction detecting circuit 3 shown in Fig.2. That is, when the permanent magnet rotor 20 is being rotated normally, the direction signal d outputted from the direction detecting circuit 3 becomes the "H" state, and when it is being rotated opposite, it becomes the "L" state. The counting circuit 4 receives the rectangular wave signal s1 outputted from the waveform shaping, circuit 2 and the direction signal d outputted from the direction detecting circuit 3 thereby to up-count or down-count the rectangular wave signal s1 in response to the direction signal d. This means that the pulse-number of the rectangular wave signal s1 generated in response to the rotation of the permanent magnet rotor 20 is up-counted or down-counted in response to the rotational direction thereof, and the amount of rotational movement of the permanent magnetic rotor 20 can be obtained from the count value of the counting circuit 4. In this case, however, the counting circuit 4 has the initial state uncertain when starting the motor and as a result, the method of providing the initial value will be shown in detail in the latter explanations on the phase coinciding operation while referring to Figs. 10 and 11. Fig. 4 shows a circuit structure of a phase detecting circuit showing an example of the circuit 10 shown in Fig. 1, and Fig. 5 shows signal waveforms of the respective member thereof. In Fig. 4, a comparator 31 has two input terminals, one connected to each end of the stator winding 11. The comparator 31 has a non-inverting input (+) connected to a terminal voltage v1 of the stator winding 11 and an inverting input (-) connected to a neutral point O of the three-phase stator windings 11, 12 and 13. The comparator 31 outputs the phase signal g to the phase adjusting circuit 5 in response to the phase voltage of the stator winding 11. In Fig. 5, a shows respective voltage waveforms of three-phase terminal voltages v1, v2 and v3 when the permanent magnet rotor 20 is being rotated, in which the terminal voltage v1 is connected to the non-inverting input (+) and the neutral point O of the three-phase stator windings 11, 12, and 13 is connected to the inverting input (-) of the comparator 31 and as a result, the comparator 31 can detect the sign of a phase voltage generated in the stator winding 11. As clear from Fig. 5 b, the rising edge of the phase signal g is coincided with the zero-crossing point of the terminal voltage v1 to the neutral point voltage O. This means that the phase signal g outputted from the phase detecting circuit 10 is to output a timing signal of the zero-crossing point of a phase voltage of one phase of the three-phase stator windings 11, 12 and 13 and as a result, the phase detecting circuit 10 can detect the phase of a phase voltage of the stator winding 11. In addition, the operation to be carried out using the phase signal g will be taken in detail in the latter explanations on the phase correction while referring to Figs. 12 and 13. First, explanations will be made on the operation of the brushless DC motor of this embodiment when it is being rotated normally. Fig. 6 is a diagram showing a signal waveform at each member of the brushless DC motor shown in Fig. 1 when it is being rotated normally, in which Fig. 6a shows respective waveforms of induce voltages e1, e2 and e3 respectively induced in the windings 11, 12 and 13, Fig. 6b shows three-phase position signals p1, p2 and p3 generated in the waveform generating circuit 6, which are generated in accordance with the rotational position of the permanent magnet rotor 20 and coincided in phase respectively with the induce voltages e1, e2 and e3, Fig. 6c shows three-phase driving currents i1, i2 and i3 supplied to the respective phases of the stator windings 11, 12 and 13, which are generated by amplifying the sine-wave position signals p1, p2 and p3 by the power supply circuit 7, and Fig. 6d shows the terminal voltages v1, v2 and v3 of respective phases of the stator windings 11, 12 and 13, which are obtained by combining the induce voltages e1, e2 and e3 induced in respective phases of the stator windings 11, 12 and 13 with the voltage drop components (only the case of the voltage v1 is shadedly shown here) due to the widing resistances generated caused by flowing the drive currents i1, i2 and i3 to the respective phases of the three-phase stator windings 11, 12 and 13. As clear from Fig. 6, the induced voltages e1, e2 and e3, position signals p1, p2 and p3, driving currents i1, i2 and i3, and terminal voltages v1, v2 and v3 are respectively brought in phase with each other and when the induced voltages e1, e2 and e3 and driving currents i1 i2 and i3 are respectively brought in phase with each other, the motor is driven at the highest efficiency. A rotating magnetic field is generated in the stator windings 11, 12 and 13 caused by the three-phase driving currents i1, i2 and i3, and due to the mutual action of the rotating magnetic field thus generated with a rotating magnetic field caused by the magnetic pole of the permanent magnet rotor 20 and the stator windings 11, 12 and 13, the permanent magnet rotor 20 is applied with a rotational force thereby to be started to rotate. Fig. 7 is a vector diagram showing a phase relation of a magnetic pole of the permanent magnet rotor 20 and a rotating magnetic field generated by the stator windings 11, 12 and 13. In Fig. 7, Φ is a magnetic pole vector showing a magnetic pole of the permanent magnet rotor 20, I is a magnetomotive force vector showing a rotating magnetic field generated by the stator windings 11, 12 and 13, and E is an induce voltage vector showing a induce voltage induced in the stator windings 11, 12 and 13. Fig. 7a shows in terms of vector the case when the motor is being rotated in the positive direction (clockwise direction) and Fig. 7b shows in terms of vector the case when it is being rotated in the opposite direction (counterclockwise direction), in each of which the magnetomotive force vector I and the magnetic pole vector Φ are rotated in the direction shown by the arrows, respectively. As clear from Fig. 7, in order to rotate the permanent magnet rotor 20 continuously, the phase of the magnetomotive force vector I generated by the stator windings 11, 12 and 13 may be always advanced by 90° in the rotational direction from the phase of the magnetic pole vector Φ of the permanent magnet rotor 20. That is, in order to rotate the motor in the normal direction, the magnetomotive force vector I may be advanced by 90° clockwise and in order to rotate it in the opposite direction, the magnetomotive force vector I may be advanced by 90° counterclockwise. As clear from Fig. 7, the magnetomotive force vector I and the induce voltage vector E are brought in phase with each other independently of the rotational direction of the motor. Under such a state, the motor is driven at the highest efficiency. Each member of the brushless DC motor of this invention for performing signal processings as shown above will be further explained in detail. Fig. 8 is a circuit diagram showing an example of an arrangement of the phase adjusting circuit 5, the waveform generating circuit 6, the initial position detecting circuit 8 and the selecting circuit 9 shown in Fig. 1. In this embodiment, the phase regulating circuit 5, waveform generating circuit 6, initial position detecting circuit 8 and selecting circuit 9 are made of a computing circuit 61, a memory 62 and digital 1 analogy converters 63, 64 and 65. The computing circuit 61 operates in conformity with a predetermined built-in program stored in a read only memory (ROM) region of the memory 62, which will be shown in detail latter, to fetch the direction demmand r inputted to the terminal 14, the phase coinciding command t inputted to the terminal 15, the phase signal g outputted from the phase detecting circuit 10 and count value c of the counting circuit 4 into a random access memory (RAM) region of the memory 62 and carries out a predetermined computing process thereby to obtain an address f. Next, the computing circuit 61 refers in response to the address f thus obtained to a sine-wave function table for one period stored in the ROM region of the memory 62 in advance thereby to obtain three-phase digital position signals dp1, dp2 and dp3 which are in response to the address f and outputs them respectively to the digital 1 analog converters 63, 64 and 65. The digital 1 analogy converters 63, 64 and 65 respectively convert the three-phase digital position signals dp1, dp2 and dp3 into analog values thereby to output the three-phase position signals p1, p2 and p3. Next, the built-in program stored in the ROM region of the memory 62 will be explained below. First, the normal mode of performing processes when rotating normally will be explained while referring to the basic flow diagram shown in Fig. 9. In Fig. 9, in a process 71, an interruption when the count value c of the counting circuit 4 is varied is waited. If the interruption is made, the process goes to a process 72. In the process 72, the count value c of the counting circuit 4, the direction command signal r inputted to the terminal 14 are taken into the RAM region of the memory 62 for storage. In a process 73, it is judged whether the direction command signal r is a clockwise one or counterclockwise one. If it is judged to be clockwise, the process goes to a process 74 in which a predetermined value (equivalent to 90° in terms of phase) is added to the count value c of the counting circuit 4 to obtain the address f. If it is judged to be counterclockwise, it goes to a process 75 in which a predetermined value (equivalent to 90° in terms of phase) is subtracted from the count value c of the counting circuit 4 to obtain the address f. The processes 71, 72, 73, 74 and 75 perform the operational processes that the phase adjusting circuit 5 does. In a process 76, three-phase addresses f1, f2 and f3 which are necessary in the next process 76 are obtained in accordance with the address signal f obtained by the process 74 or process 75. That is, the phases of the position signals p1, p2 and p3 are shifted by 120° to each other (see Fig. 6) and as a result, the three-phase addresses f1, f2 and f3 are computed as follows; f1 = f f2 = f + (120) f3 = f - (120) In addition, (120) is an address count value equivalent to 120° in terms of phase. In the process 77, the sine-wave function table stored in the ROM region of the memory 62 is referred on the basis of the address values f1, f2 and f3 obtained in the process 76 to obtain the three-phase digital potion signals dp1, dp2 and dp3. In a process 78, the three-phase digital position signals dp1, dp2 and dp3 obtained in the process 77 are outputted respectively to the digital 1 analog converters 63, 64 and 65. The digital 1 analog converters 63, 64 and 65 convert the digital position signals dp1, dp2 and dp3 into respective analog values to output the position signals p1, p2 and p3 as shown in Fig. 6. The processes 76, 77 and 78 perform the operational processes that the waveform generating circuit 6 does. After performing the process 78, it goes to the process 71 thereby to repeat the above-mentioned processes. By performing such processes by the phase adjusting circuit 5 and waveform generating circuit 6, the position signals p1, p2 and p3 are supplied to the power supply circuit 7 in response to the rotation of the permanent magnet rotor 20. The power supply circuit 7 supplies the sine-wave driving currents i1, i2 and i3 respectively to the stator windings 11, 12 and 13. That is, the amount of rotational movement of the permanent magnet rotor 20 is detected, and the magnetic field generated by the stator windings 11, 12 and 13 is rotated by the amount thus detected. As a result, the stator windings generate a rotating magnetic field, and the magnetomotive force vector I of the rotating magnetic field is formed so as to be always different in phase by 90° from the magnetic pole vector Φ of the permanent magnet rotor 20 as shown in Fig. 7. Thus, due to the mutual action of the magnetic pole vector Φ and the magnetomotive force vector I , the permanent magnet rotor 20 is applied with a rotational force thereby to continue the rotation. However, under such an initial stage that a power source is turned ON or the like, the count value of the counting circuit 4 is uncertain, resulting in a necessity to give an initial value cs of the count value thereto. Next, the operation of phase coinciding to give the initial value to the counting circuit 4 in the brushless DC motor of this embodiment will be explained in detail. When starting, the phase coinciding command signal t is inputted to the terminal 15 of the selecting circuit 9 shown in Fig. 1, and the output signal b of the initial position detecting circuit 8 is selected through the selecting circuit 9 to be outputted to the waveform generating circuit 6 as the address f. The initial position detecting circuit 8 rotates forcibly the rotating magnetic field generated by the stator windings 11, 12 and 13 clockwise or counterclockwise thereby to detect the phase relation of the magnetic pole of the permanent magnet rotor 20 and the rotating magnetic field generated by the stator windings 11, 12 and 13. Fig. 10 is a vector diagram showing a relation of the magnetic pole vector Φ generated by the permanent magnet rotor 20 and the magnetomotive force vector I generated by the stator windings 11, 12 and 13, which are for the sake of explaining the phase coinciding operation of this embodiment. When the phases are to be coincided with each other, the selecting circuit 9 outputs the address command b of the initial position detecting circuit 8 to the waveform generating circuit 6, and the address f inputted to the waveform generating circuit 6 is forcibly changed independently of the rotation of the permanent magnet rotor 20, whereby the direction of the magnetomotive force vector I of the stator windings is changed in response to the address f thus changed. As a result, by the mutual action of the magnetomotive force vector I and the magnetic pole vector Φ, the permanent magnet rotor 20 is applied with a rotational force to be started to rotate. If the motor is supposed to be unloaded, the magnetomotlve force vector I and the magnetic pole vector Φ are perfectly coincided with each other (in this case, the torque that the motor generates becomes zero), the state of which is shown in Fig. 10a. However, if the motor is supposed to be loaded, the magnetomotive force vector I and the magnetic pole vector Φ are not coincided with each other, thus making a phase angle  therebetween in accordance with the magnitude of the load. Furthermore, the phase shift direction is that when the motor is rotated clockwise, the magnetic pole vector Φ is shifted by an angle  counterclockwise from the magnetomotive force vector I as shown in Fig. 10b, and when the motor is rotated counterclockwise, the magnetic pole vector Φ is shifted by the angle  clockwise from the magnetomotive force vector I as shown in Fig. 10c. In this case, however, the counting circuit 4 up-counts or down-counts the rectangular wave signal s1 in response to the direction signal d, so that the permanent magnet rotor 20 and the amount of its rotational movement from the initial values are always detected. Here, if the initial value of the counting circuit 4 is expressed as cs, and if the magnetomotive force vector I of the stator windings is rotated by Δf clockwise, the count value c1 of the counting circuit 4 can be obtained as; c1 = cs + Δf - h where h is a count value equivalent to the phase angle  shown in Fig. 10 in terms of phase. Also, if the magnetomotive force vector I of the stator windings is rotated by Δ f counterclockwise, the count value c2 of the counting circuit 4 can be obtained as; c2 = cs - Δf + h Hence, the initial value cs of the counting circuit 4 can be calculated as; cs = (c1 + c2) / 2 Further detailed explanations will be made on the operation of performing the phase coinciding process of this embodiment while referring to Fig. 11 which is a flow diagram for explaining the phase coinciding mode in which the phases are to be coincided with each other. In Fig. 11, in a process 91, the address signal f is increased by one, and the process goes to a process 92. In this case, the initial value of the address f when phase coinciding is made zero. In the process 92, it is judged whether the address f exceeds a specific value Δf or not. If the address f does not exceed the value Δf, it is returned to the process 91 again thereby to further increase the address f by one. Then, if the address f exceeds the value Δf, it goes to a process 93. In the process 93, the count value c of the counting circuit 4 is fetched to be stored as the first count value c1 in the RAM region of the memory 62 and then, it goes to a process 94. In the process 94, the address signal f to be inputted to the waveform generating circuit 6 is decreased by one at this time, and it goes to a process 95. In the process 95, it is judged whether the address f exceeds a specific value - Δf or not. If the address f does not exceed - Δf, it is returned to the process 94 again to further decrease the address f by one. Then, if the address f exceeds - Δf, it goes to a process 96. In the process 96, the count value c of the counting circuit 4 is fetched to be stored as the second count values c2 in the RAM region of the memory 62 and then, it goes to a process 97. In the process 97, the calculation is carried out using equation (6) in terms of the first and second count values c1 and c2 obtained respectively through the processes 93 and 96 to obtain the initial value cs of the counting circuit 4. After finishing the calculating process in the process 97, it goes to a process 98. In the process 98, the value cs obtained through the process 97 is sent to the counting circuit 4 as the initial value. These processes from the process 91 to the process 98 are the operation of the phase coinciding mode in the initial state such as the case of starting the motor. Thus, the phase coinciding operation of the magnetic pole vector Φ of the permanent magnet rotor 20 and the magnetomotive force vector I generated by the stator windings 11, 12 and 13 is completed (see Fig. 10a). After completing the phase coinciding process, it goes to the normal mode shown in Fig. 9 thereby to rotate the permanent magnet rotor 20. Next, detailed explanations will be made on the phase correction to be carried out when the brushless DC motor of this embodiment is being rotated normally. Fig. 12 is a signal waveform diagram at each member of the brushless DC motor shown in Fig, 1 , in which the waveforms of the induced voltages e1, e2 and e3 induced respectively in the stator windings 11, 12 and 13, the waveforms of the voltages and currents of respective members when the phases of the driving currents i1, i2 and i3 conducted to respective phases of the stator windings 11, 12 and 13 are shifted, and the waveform of the phase signal g outputted from the phase detecting circuit 10 are shown. Fig. 12a shows the induce voltages e1, e2 and e3 induced respectively in the stator windings 11, 12 and 13, Fig. 12b shows the sine-wave position signals p1, p2 and p3 outputted from the waveform generating circuit 6 and Fig. 12c shows the driving currents i1, i2 and i3 conducted respectively to the stator windings 11, 12 and 13. The position signals p1, p2 and p3, each of which is a sine-wave signal, are sent to the power supply circuit 7 in which these sine-wave position signals p1, p2 and p3 are amplified and converted respectively into the sine-wave three-phase driving currents i1, i2 and i3 thereby to send them to respective phases of the stator windings 11, 12 and 13 and as a results, the position signals p1, p2 and p3 and the driving signals i1, i2 and i3 are respectively coincided in phase with each other. Fig. 12d shows the terminal voltages v1, v2 and v3 of respective phases of the stator windings 11, 12 and 13, which are obtained by combining the induced voltages e1, e2 and e3 induced in respective phases of the stator windings 11, 12 and 13 with the voltage drop components (only the case of the induce voltage v1 is shadedly shown here) due to the winding resistances generated caused by flowing the driving currents i1, i2 and i3 to the respective phases of the stator windings 11, 12 and 13. As clear from Fig. 12, the induce voltages e1, e2 and e3, driving currents i1, i2 and i3 and terminal voltages v1, v2 and v3 are all different in phase. Particularly, the induce voltages e1, e2 and e3 and driving currents i1, i2 and i3 are respectively shifted in phase, so that the motor is not driven at the highest efficiency. Fig. 12e shows the phases signal g outputted from the phase detecting circuit 10 shown in Fig. 4. Similar to the case shown in Fig. 5, the rising edge of the position signal g is coincided with the zero crossing point at which the terminal voltage v1 crosses the neutral point potential O toward from the downside to the upside direction, and the falling edge of the position signal g is coincided with the zero crossing point at which the terminal voltage v1 crosses the neutral point potential O toward from the upside to the downside direction. In addition, the rising edge of the phase signal g is shifted in phase by Δg from the zero crossing point of the position signal p1 (or the driving signal i1). Next, detailed explanations will be made on the phase correction to be performed in this embodiment of this invention, in which the induce voltages e1, e2 and e3 and the driving currents i1, i2 and i3 are coincided in phase with each other in response to the phase signal g outputted from the phase detecting circuit 10 as already shown in Fig. 6. Fig. 13 is a flow diagram for explaining the phase correction mode where the phase correction is carried out of this embodiment while referring to Fig. 13. In Fig. 13, in a process 131, an interruption due to the generation of the rising edge of the phase signal g outputted from the phase detecting circuit 10 is waited. If the rising edge of the phase signal g is not generated, the process goes to the normal mode operation, and if the rising edge of the phase signal g is generated, it goes to a process 132. In the process 132, the difference Δg (= f - fp) between the address value f inputted to the waveform generating circuit 6 at the time point when the rising edge of the phase signal g is generated and an address value fp which is equivalent to the zero crossing point of the position signal p1 outputted from the waveform generating circuit 6 is calculated and stored as the phase difference Δg into the RAM region of the memory 62 and then, it goes to a process 133. In the process 133, it is judged whether the magnitude of the phase difference Δg obtained through the process 132 is within the range of a predetermined value G or not. If the magnitude of the phase difference Δg is smaller than the predetermined value G, the process is shifted to the normal mode, and if the magnitude of the phase difference Δg is larger than the predetermined value G, it goes to a process 134. In the process 134, the sign of the phase difference Δg thus obtained is judged. If the sign of the phase difference Δg is positive, it goes to a process 135. In the process 135, the address value f is increased by one and then, it goes to the normal mode. If the sign of the phase difference Δg is not positive, it goes to a process 136, in which the address value f is decreased by one and then, the process is shifted to the normal mode. These processes from the process 131 to the process 136 are the operation of the phase correction mode of this embodiment. By performing the processes shown in the flow diagram of Fig. 13, the zero crossing point of each of the terminal voltages v1, v2 and v3 to the neutral point potential O is processed so as to be coincided with the corresponding one of the position signals p1, p2 and p3 outputted from the waveform generating circuit 6. This means that if the zero crossing point of the terminal voltage v1 is advanced from the address value fp corresponding to the zero crossing point of the position signal p1 as shown in Fig. 12e, the phases of the position signals p1, p2 and p3 are advanced by decreasing the address value f of the position signals inputted to the waveform generating circuit 6. On the other hand, if the zero crossing point of the terminal voltage v1 is delayed from the address value fp corresponding to the zero crossing point of the position signal p1, the phases of the position signals p1, p2 and p3 are delayed by increasing the address value f of the position signals inputted to the waveform generating circuit 6. As a result, the phases of the position signals p1, p2 and p3 (which are equal to the phases of the driving currents i1, i2 and i3) outputted from the waveform generating circuit 6 are coincided with the phases of the terminal voltages v1, v2 and v3, respectively, and the phases of the induced voltages v1, v2 and v3 are coincided with the phases of the driving currents i1, i2 and i3, respectively, as shown in Fig. 6, thus the motor being driven at the highest efficiency. Accordingly, even if the initial position cannot be precisely detected in the phase coinciding mode shown in Fig. 11, the induce voltages e1, e2 and e3 and the driving currents i1, i2 and i3 can be coincided in phase respectively with each other through the phase correction mode shown in Fig. 13 when rotating normally. In addition, even if a noise is additionally imposed on the frequency generator 1 for any reason when the motor is being driven and the count value of the counting circuit is made erroneous, the induce voltages e1, e2 and e3 and the driving currents i1, i2 and i3 can be automatically coincided in phase respectively with each other when rotating normally through the phase correction mode shown in Fig. 13. As a result, such a worst condition that the motor itself is stopped due to the accumulation of errors of the count value of the counting circuit 4 caused by the noise thus additionally imposed on the frequency generator 1 can be prevented from being developed. Fig. 14 is a circuit diagram of a phase detecting circuit showing another example of the circuit 10 shown in Fig. 4, and Fig. 15 is a diagram showing a signal waveform of each member of the phase detecting circuit shown in Fig. 14. In Fig. 14, a comparator 100 receives two terminal voltages (v2 and v3 in this case) out of the three-phase terminal voltages of the stator windings 11, 12 and 13. The comparator 100 compares the magnitudes of the two terminal voltages v2 and v3 and outputs the phase signal g to the phase adjusting circuit 5 in accordance with the result thus obtained by comparison. Fig. 15a shows a waveform of each of the three-phase terminal voltages v1, v2 and v3 when the permanent magnet rotor 20 is being rotated. The terminal voltage v2 is connected to a non-inverting input (+) of the comparator 100 and the terminal voltage v3 is connected to an inverting input (-) of the comparator 100, and if v2 > v3, the phase signal g outputted form the comparator 100 becomes "H" state, and if v2 < v3, the output g outputted from the comparator 100 becomes "L" state, which are shown in Fig. 15b. As clear from Fig. 15b, the terminal voltages v1, v2 and v3 are three-phase symmetrical signals, so that the rising edge of the phase signal g is coincided with the maximum peak point of the terminal voltage v1, and the falling edge of the phase signal g is coincided with the minimum peak point of the terminal voltage v1. That is, the phase signal g outputted from the phase detecting circuit 10 is to output a timing of the peak point of the terminal voltage v1 of the three-phase terminal voltages v1, v2 and v3 of the stator windings 11, 12 and 13, thus being capable of detecting the phase of the terminal voltage v1 by the phase detecting circuit 10. Next, detailed explanations will be made on the phase correction operation to be carried out when rotating normally in the case of using the phase detecting circuit shown in Fig. 14 of this embodiment. Fig. 16 shows the waveforms of the induce voltages e1, e2 and e3 induced respectively in the stator windings 11, 12 and 13, the waveforms of voltages and currents outputted from respective circuits when the driving currents i1, i2 and i3 conducted to respective phases of the stator windings 11, 12 and 13 are shifted in phase to each other, and the phase signal g outputted from the phase detecting circuit 10. Fig. 16a shows the induce voltages e1, e2 and e3 induced respectively in the stator windings 11, 12 and 13, Fig. 16b shows the sine-wave position signals p1, p2 and p3 outputted from the waveform generating circuit 6, and Fig. 6c shows the driving currents i1, i2 and i3 conducted respectively to the stator windings 11, 12 and 13. The position signals p1, p2 and p3 are sine-wave signals. The power supply circuit 7 amplifies these position signals p1, p2 and p3 and converts them into the sine-wave three-phase driving currents i1, i2 and i3 to supply to the phases of the stator windings 11, 12 and 13, respectively, so that the position signals p1, p2 and p3 are made equal in phase respectively to the driving currents i1, i2 and i3. Fig. 16d shows the terminal voltages v1, v2 and v3 of respective phases of the stator windings 11, 12 and 13, which are obtained by combining the induced voltages e1, e2 and e3 induced in respective phases of the stator windings 11, 12 and 13 with the voltage drop components (only the case of the induced voltage v1 is shadedly shown here) due to the winding resistances generated caused by flowing the driving currents i1, i2 and i3 to the prospective phases of the stator windings 11, 12 and 13. As clear from Fig. 16, the induce voltages e1, e2 and e3, driving currents i1, i2 and i3 and terminal voltages are all different in phase. particularly, the induced voltages e1, e2 and e3 and driving currents i1, i2 and i3 are respectively shifted in phase to each other, so that the motor cannot be driven at the highest efficiency. Fig. 16e shows the phase signal g outputted from the phase detecting circuit 10 shown in Fig. 14. Similar to the case shown in Fig. 15, the rising edge of the phase signal g is coincided with the maximum peak point of the terminal voltage v1 and the falling edge of the position signal g is coincided with the minimum peak point of the terminal voltage v1. In addition, the rising edge of the phase signal g is shifted in phase by Δg from the maximum peak point of the position signal p1 (or the driving circuit i1). Here, the explanations on the operation of performing the phase correction so as to coincide the phases of the induced voltages e1, e2 and e3 respectively with the phases of the driving currents i1, i2 and i3 in response to the phase signal g outputted from the phase detecting circuit 10 of this embodiment will be omitted for the sake of avoiding duplication because of being the same as those shown in Fig. 13. In addition, in the case of using the phase detecting circuit 10 as shown in Fig. 4, the operation is carried out so that the zero crossing point of the terminal voltage V1 of the stator winding 11 is coincided with the zero crossing point of the position signal p1 (or the driving current i1), but in the case of using the phase detecting circuit 10 as shown in Fig. 14, the operation is carried out so that the maximum peak point of the terminal voltage v1 of the stator winding 11 is coincided with the maximum peak point of the position signal (or the driving current i1). However, these two cases are not different in operation in that the phase of the position signals p1, p2 and p3 outputted from the waveform generating circuit 6, (which are equal to the phases of the driving currents i1, i2 and i3), and the phases of the terminal voltages v1, v2 and v3 are processed so as to be coincided with each other and the phases of the induce voltages e1, e2 and e3 and the phases of the driving currents i1, i2 and i3 are processed so as to be coincided with each other, respectively. In the above explanations, the power supply circuit 7 is so-called current control type power supply circuit which generates three-phase driving currents proportional respectively to the position signals, p1, p2 and p3 outputted from the waveform generating circuit 6. However, such a current amplifier that generates currents proportional respectively to the inputted position signals p1, p2 and p3 may be easy to develop an inter-phase variation in general, thus being disadvantage in that the circuit structure becomes complex as compared with a simple voltage amplifier. In the case of using the so-called voltage control type power supply circuit which generates three-phases driving voltages proportional respectively to the inputted position signals p1, p2 and p3, a phase detecting circuit as shown in Fig. 17 may be employed in stead of the phase detecting circuit 10 shown in Fig. 4. Fig. 17 shows further another example of the phase detecting circuit 10 shown in Fig. 1, and Fig. 18 shows a signal waveform of each member of the circuit shown in Fig. 17. In Fig. 17, 41 is a resistor for current detection connected in series to one-phase stator winding 11, which converts the phase current i1 conducted to the stator winding 11 into a voltage, and 42 is a comparator having two input terminals, one connected to each end of the resistor 41. The comparator 42 outputs the phase signal g to the phase adjusting circuit 5 in response to the direction of a current flowed to the resistor 41. In Fig. 18, a shows a waveform of each of the currents i1, i2 and i3 flowed respectively to three-phase stator windings 11, 12 and 13 when the permanent magnet rotor 20 is being rotated. The current i1 to be flowed to the stator winding 11 is converted through the resistor 41 into a voltage in accordance with the current itself, and the voltages at the both ends of the resistor 41 are connected respectively to a non-inverting input (+) and an inverting input (-) of the comparator 42, so that the comparator 42 can detect a sign of the phase current conducted to the stator winding 11. As clear from a waveform b shown in Fig. 18, the rising edge of the phase signal g is coincided with the zero crossing point of the current i1. That is, the phase signal g outputted from the phase detecting circuit 10 is to output a timing of the zero crossing point of one phase current of the stator windings 11, 12 and 13, so that the phase detecting circuit 10 makes possible to detect the current phase of the stator winding 11. Here, the explanations on the operation of performing the phase correction so as to coincide the phases of the induce currents e1, e2 and e3 respectively with the phases of the driving currents i1, i2 and i3 in response to the phase signal g outputted from the phase detecting circuit 10 of this embodiment will be omitted from the sake of avoiding duplication because of being approximately the same as those of the so-called current control type one shown in Fig. 12. Fig. 19 shows another example of the power supply circuit 7 shown in Fig. 1, which shows an example of using a voltage amplification type one. In Fig. 19, 141a, 141b and 141c are amplifiers which respectively receive from their inverting input (-) terminals the position signals p1, p2 and p3 outputted from the waveform generating circuit 6 through resistors 142a, 142b and 142c, and which receive a reference voltage 144 from respective non-inverting input (+) terminals. The outputs of the amplifiers 141a, 141b and 141c are connected respectively to the bases of NPN transistors 145a, 145b and 145c and PNP transistors 146a, 146b and 146c, the emitters of which are respectively connected in common. The collectors of the NPN transistors 145a, 145b and 145c are connected respectively to a positive power source, the PNP transistors 146a and 146c are connected through a current detecting resistor 148 to the ground (Gnd) and the PNP transistor 146b is connected through a current detecting resistor 147 to the Gnd. A signal ps2 and signal ps3 which are obtained by converting the currents into voltages are outputted respectively from a terminal 149 are terminal 150. In this case, however, the current detecting resistors 147 and 148 are made equal in resistance to each other. In addition, 143a, 143b and 143c are feedback resistors which are respectively connected to the common connecting points of the emitters of the NPN transistors and PNP transistors as well as to the inverting input (-) terminals of the amplifiers 141a, 141b and 141c. The outputs ps2 and ps3 from the terminals 149 and 150 are inputted to the phase detecting circuit 100 shown in Fig. 14. In this case, however, the output ps2 is inputted instead of the terminal voltage v2 and the output ps3 is inputted instead of the terminal voltage v3. Fig. 19 is a voltage amplification type power supply circuit, so that the position signals p1, p2 and p3 outputted from the waveform generating circuit 6 are amplified at an amplification factor determined by the ratio of the resistances of the resistors 142 and 143 and the sine-wave voltages thus amplified are applied respectively to the stator windings 11, 12 and 13. As a result, the stator windings 11, 12 and 13 are applied with the sine-wave driving currents i1, i2 and i3, respectively. Fig. 20 shows a signal waveform of each member of the brushless DC motor shown in Fig. 1 when rotating normally. In which, Fig. 20a shows the induce currents e1, e2 and e3 induced respectively in the stator windings 11, 12 and 13, Fig. 20b shows the sine-wave position signals p1, p2 and p3 outputted from the waveform generating circuit 6 and Fig. 20c shown the terminal voltages v1, v2 and v3 of the respective phases of the stator windings 11, 12 and 13. The position signals p1, p2 and p3 are sine-wave signals, and the power supply circuit 7 shown in Fig. 19 respectively amplifies the voltages of these signals to supply the sine-wave three-phase driving voltages v1, v2 and v3 to the respective phases of the stator windings 11, 12 and 13 and as a result, the position signals p1, p2 and p3 are made coincided in phase respectively with the driving voltages v1, v2 and v3. Fig. 20d shows the currents i1, i2 and i3 to be conducted to the respective phases of the stator windings 11, 12 and 13, in which such currents that are respectively proportional to the differences between the driving voltages v1, v2 and v3 to be applied and the induce voltages e1, e2 and e3 induced respectively in the stator windings 11, 12 and 13, or i = (v - e)/R (R = winding resistance) are flowed. That is, the signals obtained by combining the voltage drop components due to the winding resistances generated caused by the driving currents i1, i2 and i3 to be conducted respectively to the phases of the stator windings 11, 12 and 13 with the induce voltages e1, e2 and e3 induced respectively in the stator windings 11, 12 and 13 are made coincided in phase with the driving voltages v1, v2 and v3 shown in Fig. 20c. Fig. 20e shows the signals ps2 and ps3 shown in Fig. 19, in which the signal ps2 is a signal obtained by converting the current of the negative side (sink current) of the current i2 into a voltage, and the signal ps3 is a signal obtained by converting the combined current of respective currents of the negative sides (sink currents) of the currents i3 and i1 into a voltage. In addition, the fact that the signal ps3 uses the combined current of the respective negative sides of the currents i3 and i1 is because the two signals ps2 and ps3 to be inputted for comparison to the comparator 100 of the phase detecting circuit 10 shown in Fig. 14 do not become zero simultaneously, being effective to stabilize the operation of the comparator 100. Fig. 20f shows the phase signal g outputted from the phase detecting circuit 10 shown in Fig. 14, in which the rising edge of the phase signal g is coincided with the maximum peak of the current i1. As clear from Fig. 20, the rising edge of the phase signal g is shifted in phase by Δg from the maximum peak point of the position signal p1 (or the driving current v1). The respective signal waveforms of the circuits of the brushless DC motor of this embodiment shown in Fig. 20 are for the case of using a voltage control type one as the power supply circuit 7 in which the sine-wave voltages proportional respectively to the position signals are applied to the stator windings. However, the operation of performing the phase correction so as to coincide the phases of the induce voltages e1, e2 and e3 respectively with the phases of the driving currents i1, i2 and i3 in response to the phase signal g outputted from the phase detecting circuit 10 is approximately equal to that of the case of using the current control type one shown in Fig. 16 and as a result, the explanations thereon will be omitted here for the sake of avoiding duplication. As described above, the brushless DC motor of this invention generates three-phase position signals using the two-phase frequency signals which are different in phase from each other outputted from the frequency generator, which means that a position sensor such as the Hall sensor does not need to be used. In addition, in the waveform generating circuit 6 shown in Fig. 1, the sine-wave function digital data table for only one period is stored into the memory, and by referring to the function table by changing the address value by a different amount of phase, the digital three-phase position signals are outputted respectively to the three digital/analog converters 63, 64 and 65, but not limited thereto, and it is needless to say that one digital/analog converter is used to convert the digital values into analog values in a sequential manner, and the analog values thus obtained can be held respectively into three sample and hold circuits (not shown) thereby outputting them as their three-phase position signals. Also, the sine-wave function is a symmetrical periodic function and as a result, it is not necessary to store all of one period part in the function table, which means that only 1/2 period part or 1/4 period part may be stored in the function table and the remaining part may be processed appropriately according to the address value thereby to obtain the digital values equivalent to the three-phase position signals. This is advantageous in that the memory necessary for the function table can be made small in number. In addition, it is needless to say that only the function table for only one period is not stored in the memory, but the sine-wave functions for the three phases are stored in respective function tables and the digital values equivalent to the three-phase position signals are respectively outputted directly to the digital/analog converters 63, 64 and 65. Further in addition, it is needless to say that the motor to be used is made of a three-phase one, but not limited thereto, such a motor that has any number of phases can be used for this purpose.

Claims: A brushless DC motor comprising: a rotor (20) having a plurality of magnetic poles; plural-phase stator windings (11,12,13) mounted adjacent said rotor with a predetermined gap therebetween; a frequency generator (1) for generating plural-phase frequency signals each having a frequency proportional to a rotational speed of said rotor; direction detecting means (3) for detecting a rotational direction of said rotor from said plural-phase frequency signals to obtain a direction signal; counting means (4) responsive to said direction signal for up-counting or down-counting pulses of at least one frequency signal of said plural-phase frequency signals; phase adjusting means (5) for obtaining a command value from a count value of said counting means; waveform generating means (6) for generating plural-phase waveform signals in response to said command value; and power supply means (7) for supplying driving currents or driving voltages to the plural-phase stator windings in response to said plural-phase waveform signals, respectively; characterised by further comprising phase detecting means (10) for outputting a phase signal in response to a phase of a voltage or current of at least one of said plural-phase stator windings; and in that said phase adjusting means adjusts a count value of said counting means by adding thereto or subtracting therefrom a predetermined value in response to a rotational direction command and performs phase adjustment in response to said phase signal to obtain the command value. A brushless DC motor as claimed in claim 1, wherein said phase detecting means (10) compares two-phase voltages or two-phase currents to be supplied to the plural-phase stator windings to obtain said phase signal. A brushless DC motor as claimed in claim 1, wherein said phase detecting means (10) compares a current conducted to one phase of the stator windings with a sum of currents conducted to the other phases of the stator windings to obtain said phase signal. A brushless DC motor as claimed in claim 1, wherein said phase adjusting means (5) adjusts the count value of said counting means (4) in response to a phase signal only when a phase difference between a voltage and a current of said stator windings exceeds a predetermined value. A brushless DC motor as claimed in claim 1, wherein said phase adjusting means (5) adds or subtracts a predetermined value to or from the count value of said counting means (4) in response to the rotational direction command so as to rotate a phase of a rotating magnetic field generated by said stator windings by 90° in terms of electrical angle from a phase of a magnetic pole of said rotor. A brushless DC motor as claimed in claim 1, wherein said waveform generating means comprises: memory means (67) having stored therein digital data representing a sine-wave signal in advance; and digital/analog converter means (63,64,65) for converting a digital data read from said memory means into an analog value. A brushless DC motor as claimed in claim 6, wherein said waveform generating means (6) has stored therein digital data representing only one period of sine-wave signal. A brushless DC motor as claimed in claim 6, wherein said waveform generating means (6) has stored therein digital data representing only a half period or a quarter period of sine-wave signal. A brushless DC motor as claim in claim 1, wherein said counting means, phase adjusting means and waveform generating means include: memory means (62) for storing program data; and operational processing unit for performing processes in accordance with said program data. A brushless DC motor as claimed in any of claims 1 to 8 further comprising: initial position detecting means (8) for computing an initial value of said rotor from a count value of said counting means and for outputting said initial value to said counting means. A brushless DC motor as claimed in claim 10, wherein said initial position detecting means (8) detects a first count value of said counting means by rotating a magnetic field clockwise and a second count value of said counting means by rotating the magnetic field counter-clockwise and calculates an average value of said first count value and second count value as the initial value of said counting means. A brushless DC motor as claimed in claim 10, wherein said initial position detecting means (8) operates only when starting said motor. A brushless DC motor as claimed in claim 10, wherein said initial position detecting means (8) counting means, phase adjusting means and waveform generating means include: memory means (62) for storing program data; and operational processing unit means for performing processes in accordance with said program data.

Inventors: INAJI TOSHIO, UEDA EIJI, INAJI, TOSHIO, UEDA, EIJI
Applicants: MATSUSHITA ELECTRIC IND CO LTD, MATSUSHITA ELECTRIC INDUSTRIAL CO., LTD.
Application year: 1993
Main classification: H02P6
IPC codes: H02P6