MAXIMUM LIKELIHOOD SEQUENCE ESTIMATING DEVICE AND METHOD THEREFOR

Abstract: In a sampling phase calculating section, for the respective sampled value sequences y i of a received signal y(t) sampled with a plurality of different sampling phases, a complex crosscorrelation (C) between the part of the sampled value sequence y i corresponding to a training sequence Φ of an equalizer and the training sequence Φ is calculated. The sampling phase for which the absolute value of the crosscorrelation (C) is a maximum is selected, and thereafter, by the calculation based on the selected sampling phase, a sampling phase τ s for which a bit error rate BER is estimated to be a minimum is selected. In the sampling processing section (120), y(t) is sampled with τ s , and according to the sampled value sequence y, in a Viterbi algorithm processing section (130) and a transmission channel estimating section (140), an estimation value E x of a transmission symbol sequence is estimated. Thereby, the sampling phase τ s for which the bit error rate BER is estimated to be a minimum is calculated from among a plurality of different sampling phases, and a transmission symbol sequence having an improved BER is accurately estimated.

Description: The present invention relates to a maximum likelihood sequence estimator for compensating waveform distortions that occur over transmission paths, in receivers, for example in digital mobile telephony. Lately, development of digital mobile telephone systems has proceeded rapidly. However, there has been a problem that, in land digital mobile telephony, waveforms of reception signals would be considerably distorted due to a number of interference waves which bring about the delay in reception and frequency selective fading which is caused by high speed movement of mobile terminals. Therefore this distortion should be compensated by an equalizer. Maximum likelihood sequence estimation is one of the most effective equalization methods for obtaining correct transmission data from reception signal waveforms which have been distorted due to delay characteristics of the transmission paths, such as frequency selective fading. A conventional digital mobile telephone system will now be described with references to Figs. 6 and 7. The digital mobile telephone system makes use of the TDMA (Time Division Multiple Access) scheme to efficiently use limited radio frequency bands and ensure easy connection to the ISDN (Integrated Service Digital Network) services which have been implemented for the fixed telecommunications networks. An example of the frame configuration of TDMA scheme is shown in Fig. 6. In Fig. 6, one frame comprises six time slots Slot 1-6. One or two of these time slots are assigned to one subscriber. Time slots 1-6 respectively consist of a 28-bit training sequence SYNC for synchronisation and training of an equaliser, a 12-bit control information sequence SACCH, a 12-bit adjacent channel identification sequence CDVCC, data part DATA of 260 bits in total, and a 12-bit reserved area RSVD. The leading position t slot (k) of the time slot assigned to each subscriber is detected by the frame synchronisation mechanism. The position to be detected by the frame synchronisation mechanism is an estimated leading position Et slot (k) (E means estimation) to be determined by estimation. This estimated leading position Et slot (k) is merely an estimated position and therefore is not always an accurate position, and therefore it may vary depending on compensation by the frame synchronization mechanism. Fig.7 is a block diagram showing the configuration of the transmitter and the receiver for digital mobile telephony. In this transmitter/receiver configuration, the receiver 30 is connected to the output side of the transmitter 10 through the transmission line 20. The transmitter 10 is composed of an encoder 11, a transmission low-pass filter (LPF) 12, a modulator 13 and other parts. The receiver 30 is composed of a demodulator 31, a reception low-pass filter (LPF) 32, a frame synchronization mechanism 33, sampling parts 34, an equalizer 35, a decoder and other parts. Operation of the transmitter and receiver 30 is described below. The transmitter 10 converts input data bm to transmission symbols X ={x 1 , x 2 , ..., x N } through the encoder 11, controls the band width through the transmission low-pass filter, generates transmission complex base band signals s(t), modulates the transmission complex base band signals s(t) with carriers through the modulator 13 and transmits them as signals sc (t) to the transmission line 20. The receiver 30 converts signals rc (t), which has passed through the transmission line 20, to complex base band signals r (t) through the demodulator 31 and obtains reception complex base band signals y (t) which are band-limited by the reception low-pass filter 32. Then, a leading position (time) Et slot(k) of the time slot assigned to a subscriber is estimated according to these signals y(t) by the frame synchronization mechanism 33, and the reception signals y(t) are sampled by the sampling part 34 according to the following equation: where, ET slot (k): Estimated leading time of the kth assigned time slot (E means estimation.) τ 1 : Fixed sampling phase T: Symbol interval time The characteristic of the transmission line 20 due to the frequency selective fading is compensated with the sample value sequences of signals y(t) by the equalizer 35 and a transmission symbol is estimated. Finally, estimated values = {E Xn } (n=1, 2, ..., N; E means estimation) are decoded by the decoder 36 and estimated values Eb m of the transmitted data are obtained. The following describes the above described maximum likelihood sequence estimation which constitutes the equalizer. As regards the maximum likelihood sequence estimation, reference is directed to "J. G. Proakis 'Digital Communications' (1983) New York: McGraw-Hill, p. 548-554, p. 610-648". As disclosed in the above reference, the maximum likelihood sequence estimation is intended to estimate the transmission symbol sequence = {x1, x2, .., xN) for which the probability (likelihood) of materialising the reception signals sequences = {y1, y2, ..., yN) with the impulse response h(t) as known when the reception signal sequences = {y1, y2, ..., yN} in a certain range are obtained. This maximum likelihood sequence is obtained by, after all, obtaining the symbol sequence = {x1, x2, ..., xN} in which the following value is maximised if a white Gaussian noise is assumed as the noise of the transmission line. Equation (2) is efficiently calculated by using the Viterbi algorithm which is known as a decoding method for the convolutional codes. The following briefly describes the principle of the Viterbi algorithm applied to the maximum likelihood estimation while referring to Fig.8 which shows a model of the transmission path or transmission line. The transmission line is assumed in a discrete time model in which the impulse responses shown in Fig.8 are limited. In Fig.8, numerals 40 denote a delay element with a symbol interval T, 41 is a multiplier, 42 is an accumulator and 43 is an adder. |h ={hj}(j=0, ..., L) is a sample value h (t-jT) of impulse responses h(t) at the symbol interval T on the transmission line 20 including transmission and reception low-pass filters 12 and 32 and the lengths of the impulse responses are (L + 1)T. W n is the noise of the transmission line which is additive white Gaussian noise. According to this assumption, equation (2) is as given below. Therefore, the sum J n up to k=n in equation (3) is given as below using the sum J n-1 up to k=n-1. In this case, J n is a value which is proportional to the logarithmic likelihood of the reception signal sequence up to k=1 to n and referred to as the "path-metric" value and the second term of the right side of equation (4) is a value which is proportional to the logarithmic likelihood as to the state transition (branch) and referred to as the "branch-metric" value. On the other hand, the state of the transmission line model shown in Fig. 8 at timing n-1 is given by the following status vector. Sn-1 = {Xn-1, Xn-2, ..., Xn-L} If there are M types of transmission symbols, M L types of states are available for the transmission line. Assuming the transition from state S n-1 at timing n-1 to state Sn at timing n, transitions from M types of states S n-1 are available for each of M L types of states Sn. The state transition between these timings is referred to as "branch" and Mn types of paths which follow up the states are referred to as "path". There are the paths from M types of states as the preceding timing for each of M L types of available states at respective timings. The Viterbi algorithm is for calculating the path-metric value of equation (4) with respect to M number of available paths in respective states at respective timings to select a path with the largest value. Accordingly M L types of paths are left at each timing and the past paths are gradually converged to one path. An estimated value of the transmission symbol sequence is obtained from the path which is finally converged. Fig. 9 is a block diagram showing a configuration example of the conventional maximum likelihood sequence estimator. This maximum likelihood sequence estimator consists of a Viterbi algorithm processing part 50 and a transmission line estimating part 60. Reception signals y(t) are sampled according to equation (1) by the sampling parts when the kth assigned time slot is to be processed and entered into the Viterbi algorithm processing part 50 and the transmission line estimating part 60. The transmission line estimating part 60 estimates impulse responses of the transmission line according to the adaptive algorithm such as LMS (Least Mean Square) from the host equipment, not shown, which provides information necessary for the sample value sequences y and the component units in the receiver 30 shown in Fig. 7 which corresponds to the sample value sequence by using the given training sequencer shown in Fig. 7 in the training sequence part of the time slot shown in Fig. 6 of the sample value sequences . In the sample value sequences other than the training sequence part of the time slot shown in Fig. 6, the estimation of the impulse response of the transmission line is continues using the estimated value of the transmission symbol sequence. The estimated impulse responses Eh of the transmission line are entered into the Viterbi algorithm processing part. The Viterbi algorithm processing part 50 estimates a transmission symbol according to the above described principle by using the impulse response Eh of the transmission lines which have been estimated as the sample value sequence for reception signals. However, the conventional maximum likelihood sequence estimator carries out simulation under the conditions that are applied in advance and fixes the sampling phase at a trade-off point which is determined by checking the bit error rate and therefore there is a problem that an optimum sampling phase where the bit error rate is minimised under all conditions is not established. This is exemplified by GB-A-2 242 339 which discloses a circuit for estimating frame phase, which includes sampling means for sampling reception signals at a symbol interval or period (T) with respect to respective sampling phases of sampling phase sequences of the reception signals, correlation value calculating means for obtaining complex cross-correlation functions of the sample value sequences and training sequences of the sample value sequences, and selecting means for obtaining a selected sampling phase at which the absolute values of the complex cross-correlation functions are maximised, by comparing the absolute values. Reference is also directed to EP-A-0 544 315 (Art. 54(3) EPC), which discloses a maximum likelihood sequence estimation apparatus having a plurality of samplers which sample an input signal at a constant pulse frequency in sequentially different sampling phases. In addition, the conventional maximum likelihood sequence estimator has an accompanied problem that, if the number of interference waves and the delay time varies, the optimum sampling phase which minimises the bit error rate vary accordingly and moreover the deviation of the sample phase for the optimum sampling phase may cause deterioration of the bit error rate, which has proved to be a difficult problem to solve. An object of the present invention is to provide a maximum likelihood sequence estimator capable of solving the above described problems. According to the present invention, there is provided a maximum likelihood sequence estimator comprising sampling means for sampling reception signals y(t) from a transmission path, at a symbol interval T with respect to respective sampling phases of sampling phase sequences τ = {τ (i)} (i = 1, 2, ..., M), to obtain sample value sequences i = {yin} (i = 1, 2, ..., M; n = 1, 2, ..., Nt) of the reception signals y(t) correlation value calculating means for obtaining complex cross-correlation functions = {Ci} (i = 1, 2, ..., M) of the sample value sequences i and training sequences = {n} (n = 1, 2, ..., Nt) of the sample value sequences i, and selecting means for obtaining a selected sampling phase at which the absolute values of the number M of complex cross-correlation functions Ci are maximised by comparing the absolute values; a transmission path estimating part for obtaining estimated values E|h = {Ehj} (j = 0, 1, ... L,) of impulse responses of the transmission path according to an adaptive equalising algorithm with respect to the sample value sequences y and the corresponding training sequences thereto and from estimated values of the transmission symbol sequences and the sample value sequences ; and a Viterbi algorithm processing part for obtaining the estimated values = {Exn} (n = 1, 2, ..., N) of transmission symbol sequences from the sample value sequences y based on the estimated values E|h, characterised by sample phase smoothing means for obtaining a sampling phase τs based on the selected sampling phase obtained by said selecting means such that the bit error rate is minimised, a sampling processing part for sampling the reception signals y(t) entered at specified timings with respect to a symbol interval T and the sampling phase τs to obtain sample value sequences = {yn} (n = 1, 2, ..., N). According to the invention, there is also provided a maximum likelihood sequence estimating method comprising sampling reception signals y(t) from a transmission path, at a symbol interval T with respect to respective sampling phases of sampling phase sequences τ = {t (i)} (i = 1, 2, ..., M), to obtain sample value sequences i = {yin} (i = 1, 2, ..., M; n = 1, 2, ..., Nt) of the reception signals y(t) obtaining complex cross-correlation functions = {Ci} (i = 1, 2, ..., M) of the sample value sequences y i and training sequences = {n} (n = 1, 2, ..., Nt) of the sample value sequences i, and selecting a sampling phase at which the absolute values of the number M of complex cross-correlation functions Ci are maximised by comparing the absolute values, obtaining estimated values E|h = {Ehj} (j=0, 1, ... L), of impulse responses of the transmission path according to an adaptive equalising algorithm with respect to the sample value sequences and the corresponding training sequences thereto and from estimated values of the transmission symbol sequences and the sample value sequences y, and using the Viterbi algorithm to obtain the estimated values = {Exn} (n = 1, 2, ..., N) of transmission symbol sequences from the sample value sequences based on the estimated values E |h, characterised by obtaining a sampling phase τs based on the selected sampling phase such that the bit error rate is minimised, sampling the reception signals y(t) entered at specified timings with respect to a symbol interval T and the sampling phase τs to obtain sample value sequences = {yn} (n = 1,2,...,N). A maximum likelihood sequence estimator in accordance with the present invention, described in more detail hereafter, for use with a transmission line, is provided with a sampling phase calculating part, a sampling processing part, a transmission line estimating part and a Viterbi algorithm processing part. The sampling phase calculating part performs the function of sampling the reception signals y(t) from the transmission lines at the symbol interval T in respective sampling phases of the sampling phase sequence τ = {τ(i)} (i=1, 2, ..., M) by starting from the estimated leading time Et slot (k) of the time slot, obtaining the sample value sequences of M number of the above described different reception signals y(t) as shown below, (i=1, 2, ..., M; n=1, 2,..., Nt) calculating the complex cross-correlation functions ={Ci} (i=1, 2, ..., M) of the training sequences from the above sample value sequences y/ i and the training sequences , comparing the absolute values of these M complex cross-correlation functions Ci, selecting a sampling phase τ(I k ) where the absolute values are maximized and obtaining the specified sampling phase τ s from calculation based on the selected sampling phase τ(I k ). The sampling processing part performs the function of sampling the above described reception signals y(t), which are entered at the specified timing, at the sampling phase τ s and the symbol interval T, starting from the estimated leading time Et slot (k) and obtaining the sample value sequences = {y n } (n=1, 2, ..., N) . The transmission line estimating part has the function for obtaining the estimated value E |h = {Ehj} (j = 0, 1, ..., L) of impulse responses of the above transmission lines according to the adaptive equalizing algorithm with respect to the sample value sequences and the training sequences corresponding thereto and also the estimated value of the transmission symbol sequence and the sample value sequence . The Viterbi algorithm processing part has the function for obtaining the estimated values of the above described transmission symbol sequences from the sample value sequences according to the Viterbi algorithm on the basis of the above estimated value E |h. The sampling phase calculating part may calculate complex cross-correlation functions C of the training sequence of the equaliser from the portions corresponding to the sample value sequences i and the training sequences of the equaliser for the sample value sequences with respect to individual sample value sequences i for reception signals y(t) which have been sampled in a plurality of different sampling phases. After a sampling phase τ(Ik) where the complex cross-correlation function C i is maximised has been selected, a sampling phase τs where the bit error rate is minimised is calculated through calculation based on the selected sampling phase τ(Ik) and given to the sampling process part. The sampling process part sample reception signals y(t), which are entered at the specified timings, with the sampling phase τs and the symbol internal T to obtain the sample value sequence and sends it to the transmission line estimating part and the Viterbi algorithm processing part. The transmission line estimating part functions to obtain estimated values E|h of impulse responses of the transmission line from the sample value sequences and the training sequences corresponding to the sample value sequences i in the training sequence part of the sampling value sequences i and estimated values E |h of impulse responses of the transmission lines by using the estimated values Ex of the sample value sequences and the transmission symbol sequences for reception signals y(t) in the part other than the training sequences of the sampling value sequences i, and the estimated values E|h obtained are given to the Viterbi algorithm processing part. The Viterbi algorithm processing part functions to obtain the estimated values of the transmission symbol sequences Ex based on the estimated values E h of the impulse responses of the transmission lines. Thus the present invention enables calculation of the sampling phases to be made simply and accurately where the bit error rate is estimated to be least from a plurality of different sampling phases and enable estimates to be made which provide a more precise transmission symbol sequence in which the bit error rate is improved. Accordingly, the present invention also enables to solve the problem that the optimum sampling phase where the bit error rate is minimized under all conditions is not always established since the sampling phase is fixed as in the conventional case and also the problem, if the number of interference waves and the amount of delay differ, the optimum sampling phase where the bit error rate is minimised differs and moreover the deviation of the sampling phases from the optimum sampling phase causes deterioration of the bit error rate. This example of the invention will now be described in more detail with reference to the accompanying drawings, in which: Fig. 1 is a diagram showing a configuration of a maximum likelihood sequence estimator which is a first embodiment of the present invention; Fig. 2 is a flow chart showing operation of a sampling processing part of the present invention; Fig. 3 is a flow chart showing operation of a transmission line estimating part according to the present invention; Fig. 4 is a Viterbi algorithm processing part according to the present invention; Fig. 5 is a flow chart showing operation of a sampling phase calculating part according to a first embodiment of the present invention; Fig. 6 is a diagram showing a frame configuration of TDMA scheme; Fig. 7 is a diagram showing a configuration of a transmitter and a receiver for digital mobile telephony; Fig.8 is a diagram showing a model of transmission line shown in Fig.7; Fig.9 is a diagram showing a configuration of a conventional maximum likelihood sequence estimator; Fig.10 is a sampling phase characteristic diagram when the amount of delay shown in Fig.1 is 0.25T (10µ sec); Fig.11 is a sampling phase characteristic diagram when the amount of delay shown in Fig.1 is 1.00T (40µ sec) ; Fig.12 is a diagram showing a configuration of a maximum likelihood sequence estimator according to a second embodiment of the present invention; Fig.13 is a flow chart showing operation of a sampling phase calculating part according to a second embodiment of the present invention; Fig.14 is a diagram showing a configuration of a maximum likelihood sequence estimator according to a third embodiment of the present invention; Fig.15 is a flow chart showing operation of a sampling phase calculating part according to a third embodiment of the present invention; Fig.16 is a diagram showing a configuration of a maximum likelihood sequence estimator according to a fourth embodiment of the present invention; and Fig.17 is a flow chart showing operation of a sampling phase calculating part according to a fourth embodiment of the present invention. First Embodiment In the first embodiment of the present invention, a maximum likelihood sequence estimator for forming, for example, an equalizer 35 and sampling parts 34 in a receiver 30 shown in Fig.7 is described. The embodiment is described as to the case where a sampling phase τs which is estimated to provide the least bit error rate is calculated from the sequence τ = {τ (i)} (i=1, 2, ..., M) of simple non-increasing or simple non-decreasing sampling phases. Fig.1 shows a configuration of a maximum likelihood sequence estimator which is an embodiment of the present invention and Fig.1 (a) a functional block diagram of the whole maximum likelihood sequence estimator and Fig.1 (b) is a functional diagram of the sampling phase calculating part shown in Fig.1 (a). In Fig.1 (a), the maximum likelihood sequence estimator comprises a program control which employs a discrete circuit such as an integrated circuit or a processor. The maximum likelihood sequence estimator is provided with a sampling phase calculating part 100, a delay processing part 110, a sampling processing part 120, a Viterbi algorithm processing part 130 and a transmission line estimating part 140. The sampling processing part 120 is connected to the output sides of the sampling phase calculating part 100 and the delay processing part 110. The Viterbi algorithm processing part 130 and the transmission line estimating part 140 are connected to the output side of the sampling processing part 120. The sampling phase calculating part 100 has the functions for sampling reception signals y(t) from the transmission line, calculating the specified sampling phase τs through calculations for sample value sequences i = {y in } (i=1, 2, ..., M; n=1, 2, ..., Nt) and the training sequences SYNC ={n} (n=1, 2, ..., Nt) and giving the sampling phase obtained to the sampling processing part 120. The delay processing part 110 functions to give the reception signals y(t) which have been delayed as long as specified to the sampling processing part 120. The sampling processing part 120 functions to obtain the sample value sequence = {y n } (n = 1, 2, ..., N) by sampling reception signals y(t) from the delay processing part 110 with the sampling phase τs and the symbol interval T, starting from the estimated leading time Et slot (k) of the time slot when the assigned kth time slot is to be processed and gives it to the Viterbi algorithm processing part 130 and the transmission line estimating part 140. The Viterbi algorithm processing part 130 functions to obtain the estimated values = {Exn} (n = 1, 2, ..., L) of the transmission symbol sequence according to the principle of the Viterbi algorithm, which has conventionally been used, in accordance with the input of the sample value sequence and the estimated values E|h = {Eh j } (j = 0, 1, ..., L) and comprises branch metric calculation means, path metric calculation means and others. The transmission line estimating part 140 functions to obtain the estimated values E of the impulse responses of the transmission line from the sample value sequences and the training sequences corresponding thereto in the training sequence part of the time slot shown in Fig.6 of the sample value sequences and the estimated values E of the impulse responses by using the sample value sequences of reception signals y(t) and the estimated values Ex of the transmission symbol sequences in the part other than the training sequence part of the time slot shown in Fig.6 of the sample value sequences , and give the estimated values of impulse responses to the Viterbi algorithm processing part 130. Switch means 150 functions to switch over the transmission line estimating part 140 to the training sequence when processing the training sequence part of the time slot shown in Fig.6 of the sample value sequences and to the estimated value Ex side of the transmission symbol sequences when processing the part other than the training sequence part and gives the switchover signals to the transmission line estimating part 140. Fig.2 is a flow chart showing the operation of the sampling processing part, Fig.3 is a flow chart showing the operation of the Viterbi algorithm processing part and Fig.4 is a flow chart showing the operation of the transmission line estimating part. The operation of the maximum likelihood sequence estimator in accordance with the present invention is described below referring to Figs.2, 3 and 4 and Fig.1(a). Reception signals y(t) are entered into the sampling phase calculating part 100 and the delay processing part 110 for each assigned time slot. For processing the kth assigned time slot, the sampling phase calculating part 100 samples reception signals at M number of different sampling phases τ (i) and the symbol interval T, starting from the estimated leading time Et slot (k) of the time slot. Subsequently, the sampling phase calculating part 100 calculates the complex cross-correlation functions of the part corresponding to the training sequences of the equalizer as given below. {y (Et slot (k) + τ (i) + (n - 1) T} (i = 1, 2, ..., M; n = 1, 2, ..., Nt) and the training sequences of the equalizer of the sample value sequences are calculated for each i value. The sampling phase where the absolute value of this complex cross-correlation function is maximized is selected and the sampling phase τs calculated based on the calculated sampling phase is given to the sampling processing part 120. On the other hand, reception signals y(t) entered into the delay processing part 110 are delayed until the sampling phase τs is selected by the sampling phase calculating part 100 and then entered into the sampling processing part 120. The sampling processing part 120 receives the reception signals y(t) and the sampling phases τs (ST201) and, when processing the kth assigned time slot, samples reception signals y(t) according to the following equation (ST202), where, Et slot (k): Estimated leading time of the kth assigned time slot τs: Sampling phase T: Symbol interval time and the sample value sequences are given to the Viterbi algorithm processing part 130 and the transmission line estimating part 140 (ST203). The Viterbi algorithm processing part 130 functions to obtain the estimated values Ex from the sample value sequences from the sampling processing part 120 and the estimated values E (ST301) from the transmission line estimating part 140 according to the above described principle, and outputs the estimated values (ST303). The transmission line estimating part 140 functions to obtain the estimated values E of the impulse responses of the transmission lines from the sample value sequences and the corresponding training sequences (ST403) thereto which are switched over and entered by switch means 150 according to the adaptive equalizing algorithm in the training sequence SYNC part of the time slot down in Fig.6 of the received sample value sequences (ST404) and give the estimated values to the Viterbi algorithm processing part 130 (ST407). In addition, the transmission line estimating part 140 functions to obtain the estimated values E of the impulse responses of the transmission lines (ST406) by using the sample value sequences of reception signals y(t) and the estimated values Ex (ST405) of the transmission symbol sequences (which are switched over and entered from switch means 150) and give the estimated values E to the Viterbi algorithm processing part 130 (ST407) in the part other than the training sequence part of the time slot shown in Fig.6 of the sample value sequences The configuration of the sampling phase calculating part 100 is shown in Fig.1(b) which is a feature of the present invention. This sampling phase calculating part 100 is provided with sampling parts 101-1 ~ 101-M which sample reception signals y(t) at the sampling phases of the above described sampling phase sequence τ and the symbol interval T, starting from the estimated leading time Et slot (k) of the time slot and obtain M number of sample value sequences i of reception signals y(t). A maximum correlation index selector 103 is connected to the output sides of sampling parts 101-1 ~ 101-M through the correlation value calculators 102-1 ~ 102-M. The correlation value calculators 102-1 ~ 102-M function to calculate complex cross-correlation functions Ci (i=1, 2, ..., M) from the sample value sequence i and corresponding training sequences The maximum correlation index selector 103 compares the absolute values of M complex cross-correlation functions Ci which are calculated by the correlation value calculators 102-1 ~ 102-M and selects the sampling phase τ (I k ) where the absolute values are maximized. In addition, the maximum correlation index selector functions to output the index I k of the selected sampling phase τ (Ik) as the index I k of the sampling phase which is selected when the kth sampling phase is selected, and a sampling phase smoother 104 is connected to the output side thereof. The sampling phase smoother 104 functions to obtain the sectional weighted mean value, sectional maximum value and sectional minimum value of index I k selected by the maximum correlation index selector 103 and outputs the sampling phase τ s corresponding to the index which is obtained by adding a value obtained by subtracting the sectional minimum value from the sectional maximum value and multiplying it by a constant to the sectional weighted mean value. Fig.5 is a flow chart showing the operation of the sampling phase calculating part according to the first embodiment. The operation of the sampling phase calculating part 100 is described below, referring to Figs.5 and 1(b). Reception signals y(t) are entered for each assigned time slot and sampled in the sampling parts 101-1 ~ 101-M for every finite section according to the following equation when processing the kth assigned time slot and entered into the correlation value calculators 102-1 ~ 102-M (ST501). i = {y(Et slot (k) + τ(i) = (n-1) T} (i = 1, 2, ..., M; n = 1, 2, ..., Nt) The correlation value calculators 102-1 ~ 102-M calculate the complex cross-correlation functions C i of the sample value sequences i and corresponding training sequences of reception signals y(t) according to the equation (8) given below and gives these functions to the maximum correlation index selector 103 (ST502): where, * means that complex conjugation is used. The maximum correlation index selector 103 selects the sampling phase τ (I k ) where the absolute values of M complex cross-correlation functions C i are maximized and gives this index to the sampling phase smoother 104 as the index I k of the sampling phase which has been selected (ST504-ST507). where, W a (j)(j = 1, 2, ..., L s ) : Load coefficient Ls; Maximum length of calculation Is = [Iavg (k) + Ws (Imax (k) - Imin (k))] where, [ ]: Gaussian notation (The maximum integer which does not exceed a value in the notation is returned.) W s : Shift coefficient I s : Selection sampling phase index Figs.10 and 11 respectively show the results of simulation of the bit error rate in the case that this embodiment is applied. For this simulation, the frame configuration is of the North American Standard Digital Cellular Scheme shown in Fig.6 and the transmission line model is a 2-wave non-correlated Rayleigh fading model. Fig.10 is a sampling phase characteristic diagram when the amount of delay shown in Fig.1 is 0.25T (10µ sec) and Fig.11 is a sampling phase characteristic diagram when the amount of delay shown in Fig.1 is 1.00T (40µ sec). The horizontal axes of Figs.10 and 11 denote the sampling phase deviation from the center of roll-off wave of a direct wave and the vertical axes denote the bit error rate (BER). From Figs.10 and 11, it is known that the bit error rate in the maximum likelihood sequence estimator of this embodiment when the sampling phase deviates from the optimum point is smaller than that in case of the conventional maximum likelihood sequence estimator. Thus, in this embodiment, complex cross-correlation functions Ci for the training sequences of the equalizer of the parts {y (Et slot (k) + τ (i) + (n-1)T} (n = 1, 2, ..., Nt) corresponding to the training sequences of the sample value sequences i of reception signals y(t) are calculated by the correlation value calculators 102-1 ~ 102-M with respect to each of a plurality of different sampling phases τ (i) when the reception signals are sampled with the above described sampling phases. Samples phases τ s are calculated from the sampling τ (i) where the complex cross-correlation function C i are maximized by the maximum correlation index selector 103 and the sampling phase smoother 104. Therefore the bit error rate (BER) can be improved better than in the embodiments according to the prior art. Second Embodiment In the second embodiment, there is described a maximum likelihood sequence estimator for forming, for example, the equalizer 35 and the sampling parts 34 in the receiver 30 shown in Fig.7. The following describes an embodiment when the sampling phase τs where the bit error rate is minimized is calculated from simple non-increasing or simple non-decreasing sampling phase sequences τ = {τ (i)} (i = 1, 2, ..., M). Fig.12 shows a configuration of a maximum likelihood sequence estimator which is the second embodiment of the present invention. Fig.12 (a) shows a functional block diagram of the whole sampling phase calculating part and Fig.12(b) is a functional block diagram of the sampling phase calculating part shown in Fig.12(a). In this case, the same components as those of the first embodiment are given the same symbols thereof. The second embodiment differs from the first embodiment only in the configuration of the sampling phase smoother in the sampling phase calculating part. The configuration of the sampling phase smoother 201 in the sampling phase calculating part 200 which is the second embodiment is described below, referring to Fig.12(b). The sampling phase smoother 201 functions to obtain the weighted mean value, a maximum value and a minimum value of the index I k selected by the maximum correlation index selector 103 and outputting a sampling phase τ s corresponding to the index I k obtained by subtracting the minimum value from the maximum value, multiplying it by an α constant and adding the value obtained to the weighted mean value. Other elements of the second embodiment shown in Fig.12 are the same as that of the first embodiment and therefore the description is omitted. Fig.13 is a flow chart showing the operation of the sampling phase calculating part of the second embodiment according to the present invention. The operation of the sampling phase calculating part 200 is described below, referring to Fig.12 (b) and Fig.13. The sampling phase calculator 200 according to the second embodiment, as in the first embodiment, obtains complex cross-correlation functions Ci (i = 1, 2, ..., M) from reception signals y(t) and training sequences through the sampling parts 101-1 ~ 101-M and the correlation value calculating parts 102-1 ~ 102-M and gives these functions to the maximum correlation index selector 103 (ST1301, ST1302). When the index I k is supplied from the maximum correlation index selector 103 to the sampling phase smoother 201 (S1303), the sampling phase smoother 201 outputs the sampling phase τs obtained from the above described equations (12) and (13) and the following equations (14) ~ (17) at the time of selection of the kth sampling phase (ST1304 ~ ST1307). I MAX (k) = max I j 1 ≤ j ≤ k I MIN (k) = max I j 1 ≤ j ≤ k I AVG (k) = (1- α) I AVG (k-1) + I k where, α): constant I AVG (o) = 0 The second embodiment described above enables to provide the almost same effects as the first embodiment and speed up the estimation processing because a volume of calculation in the sampling phase smoother 201 is small. Third Embodiment The third embodiment describes a maximum likelihood sequence estimator for forming the equalizer 35 and the sampling part 34 in, for example, the receiver 30 shown in Fig.7. The embodiment is described with respect to a case for calculating a sampling phase τ s where the bit error rate is estimated to be minimum from an optional sampling phase sequence τ = {τ (i)} (i = 1, 2. ..., M). Fig.14 shows a configuration of a maximum likelihood sequence estimator which is the third embodiment of the present invention and Fig.14 (a) is a functional block diagram of the maximum likelihood sequence estimator and Fig.14 (b) is a functional block diagram of the sampling phase calculating part of the sampling phase calculating part shown in Fig.14 (a). In this case, the same components of the first embodiment are given the same symbols. In the third embodiment, the internal configuration of the sampling phase calculating part differs from that of the first embodiment. The following describes the configuration of the sampling phase calculating part 300, referring to Fig.14 (b). In Fig.14 (a), the sampling phase calculating part 300 is provided with sampling parts 101-1~101-M which sample reception signals y (t) at respective sampling phases of the sampling phase sequences τ = {τ (i)} (i = 1, 2, ..., M) and the symbol interval T as the symbol time, starting from the estimated leading time Et slot (t) of the time slot and obtain M number of different sample value sequences i = {y in } (i = 1, 2, ..., M; n = 1, 2, ..., Nt) of reception signals. The maximum correlation sampling phase selector 302 is connected to the output sides of sampling parts 101-1 ~ 101-M through the correlation value calculators 102-1 ~ 102-M. The correlation value calculators 102-1 ~ 102-M function to calculate complex cross-correlation functions C i (i = 1, 2, ..., M) of the sample value sequences i and the training sequence SYNC = { n} (n = 1, 2, ..., Nt). The maximum correlation sampling phase selector 302 functions to compare the absolute values of M number of complex cross-correlation functions C i calculated by the correlation value calculators 102-1~102-M, select a sampling phase τ sel (k) where the absolute values of the correlation value functions are maximized and output the selected sampling phase, and the sampling phase smoother 301 is connected to the output sides of the correlation value calculators. The sampling phase smoother 301 functions to obtain a coordinate transformation term tc (k) for transforming the sampling phase to the coordinates in reference to the input timing of reception signals when the frame synchronization is established in accordance with the estimated leading time Et slot (k) from the frame synchronization mechanism. The sampling phase smoother 301 functions to obtain the sectional weighted mean value, the sectional maximum value and the sectional minimum value after adding the coordinate transformation term tc (k) to the sampling phase τ sel (k) selected by the maximum correlation sampling phase selector 302, add a value obtained by subtracting the absolute values of the sectional minimum value and the coordinate transformation term tc (k) from the sectional maximum value and multiplying it by a constant to the sectional weighted mean value and output the sampling phase τ s obtained by subtracting the coordinate transformation term tc (k) from the resultant value.. Another configuration of the third embodiment shown in Fig.14 is the same as that of the first embodiment and therefore the description is omitted. Fig.15 is a flow chart showing the operation of the sampling phase calculating part according to the third embodiment of the present invention. The operation of the sampling phase calculating part 300 is described below, referring to Figs.15 and 14(b). Reception signals are entered for each assigned time slot and sampled by the sampling parts 101-1 ~ 101-M according to equation (7) when the kth assigned slot is processed, then entered respectively into the correlation value calculators 102-1~102-M (ST1501). Each of correlation value calculators 102-1 ~ 102-M calculates complex cross-correlation functions C i of the sample value sequences i and corresponding training sequences of reception signals y(t) according to equation (8) and gives the functions obtained to the maximum correlation sampling phase selector 302 (ST1502). The maximum correlation sampling phase selector 302 selects a sampling phase τ sel (k) where the absolute values of M number of cross-correlation functions C i are maximized and gives the selected sampling phase to the sampling phase smoother 301 as the sampling phase τ sel (k) when the kth sampling phase is selected (ST1504). The sampling phase smoother 301 outputs the sampling phases τ s obtained according to the following equations (18)~(23) (ST1504~ST1507). Δ tc (k) = Et slot (k) - Et slot (k-1) k > 1 τ s = τ avg(k) - tc (k) + W s (τ max(k)) - τ min(k) - | tc (k) | ) where, W s : Shift coefficient W a (j) (j-1, 2, ..., L s ): Weighted coefficient L s : Maximum calculated length Fourth Embodiment In the fourth embodiment, there is described a maximum likelihood sequence estimator for forming the equalizer 35 and sampling parts 34 in a receiver 30 as shown, for example, in Fig.7. In this case, the embodiment is described with respect to the case for calculating a sampling phase τ s , where the bit error rate is minimized, from optional sampling phase sequences τ={ τ (i) } (i = 1, 2, ..., M). Fig.16 shows a configuration of the maximum likelihood sequence estimator which is the fourth embodiment according to the present invention, and Fig.16 (a) is a functional block diagram of the whole maximum likelihood sequence estimator and Fig.16 (b) is a functional block diagram of a sampling phase calculating part shown in Fig.16 (a). The same components as those of the third embodiment are given the same symbols. The fourth embodiment differs from the third embodiment in that the configuration of the sampling phase smoother in the sampling phase calculating part differs from that of the third embodiment. The configuration of the sampling phase smoother 401 in the sampling phase calculating part 400 of the fourth embodiment is described below, referring to Fig.16 (b). The sampling phase smoother 401 functions to obtain a first coordinate transformation term t c1 (k) and a second coordinate transformation term t c2 (i, k) for transforming the sampling phase to the coordinates in reference to the input timing of reception signals in frame synchronization at a current time in accordance with the estimated leading time ET slot (k) from the frame synchronization mechanism 33 (refer to Fig.7). The sampling phase smoother 401 functions to obtain the sectional weighted mean value, the sectional maximum value and the sectional minimum value after adding the second coordinate transformation term t c2 (j, k) to the sampling phase τ sel (k) selected by the maximum correlation sampling phase selector 302 and output the sampling phase τs obtained by adding a value, which is obtained by subtracting the absolute values of the sectional minimum value and the first coordinate transformation term t c1 (k) from the sectional maximum value and multiplying it by a constant, to the sectional weighted mean value. Another configuration of the fourth embodiment shown in Fig.16 is the same as that of the third embodiment and therefore the description is omitted. Fig.17 is a flow chart showing the operation of the sampling phase calculating part according to the fourth embodiment of the present invention. The operation of the sampling phase calculating part 400 is described below, referring to Figs.17 and 16(b). The sampling phase calculating part 400 of the fourth embodiment, as in the first embodiment, functions to obtain complex cross-correlation functions = {Ci} (i = 1, 2, ..., M) from the reception signals y(t) and training sequences by the sampling parts 101-1 ~ 101-M and the correlation value calculators 102-1 ~ 102-M and give these functions to the maximum correlation sampling phase selector 302 (ST1701, ST1702). When the maximum correlation sampling phase selector 302 selects a sampling phase τ sel (k) where the absolute values of correlation functions C i are maximized and gives this selected sampling phase to the sampling phase smoother 401 as the sampling phase τ sel (k) which is selected in the kth selection of the sampling phase (ST1703), the sampling-phase smoother 401 outputs the sampling phase τ s obtained according to the following equations (24) ~ (30) in the kth calculation of the sampling phase (ST1704~ST1707). τs = τAVG (k) + ws(τMAX (k) - τMIN (k) - |t c1 (k)|) where, W s : Shift coefficient W a : (j) (j = 1, 2, ..., L s ); Weighted coefficient L s : Maximum calculation length The reference point of the sampling phase selected in respective time slots according to timing compensation by the frame synchronization mechanism differs with each time slot. In calculation in the first embodiment for which the above point is not taken into account, the bit error rate in calculation for obtaining the sampling phase τ s may be adversely affected but, in the third and fourth embodiments, an adverse effect on calculation for obtaining the sampling phase τ s can be prevented. The present invention is not limited to the above described embodiments and is available in various modifications. Examples of such modifications are as shown below. (i) A matching filter and a matching filter for whitening can be added to the configurations as the maximum likelihood sequence estimator shown in, for example, Figs.1, 12, 14 and 16 and the output of this whitening filter can be entered into the Viterbi algorithm processing part 130. These configurations differ only in the arithmetic equations for path-metric values and therefore the above described embodiments can be similarly applied to the maximum likelihood sequence estimators with those configurations. (ii) The sampling phase selecting parts described in respective embodiments can be composed of the circuits other than shown in Figs.1 (b), 12 (b), 14 (b) and 16 (b). In addition, the embodiments are adapted to obtain the estimated values of transmission symbols by using the TDMA frame configuration shown in Fig.6 but the above embodiments can be applied using other data sequences. The embodiments of the present invention described above are for use in land mobile communications equipment such as cordless telephone systems, automobile telephone systems, portable telephone systems, pocket type bells, simple land radio telephone systems, teleterminal systems, etc., maritime mobile communications equipment such as maritime mobile radio telephone systems, maritime satellite communications systems or aeronautical mobile communications systems such as airborne public telephone, etc. In addition, the present invention is also adapted for use in radio mobile telephone systems which are used by various public organizations, MCA land mobile radio communications systems, self-defense communications systems for taxi radio communications, specific small power radio stations and others.

Claims: A maximum likelihood sequence estimator comprising: sampling means (101-1, 101-M) for sampling reception signals y(t) from a transmission path, at a symbol interval T with respect to respective sampling phases of sampling phase sequences τ = {τ (i)} (i = 1, 2, ..., M), to obtain sample value sequences i = {yin} (i = 1, 2, ..., M; n = 1, 2, ..., Nt) of the reception signals y(t) correlation value calculating means (102-1, 102M) for obtaining complex cross-correlation functions = {Ci} (i = 1, 2, ..., M) of the sample value sequences i and training sequences = {n} (n - 1, 2, ..., Nt) of the sample value sequences i, and selecting means (103, 302) for obtaining a selected sampling phase at which the absolute values of the number M of complex cross-correlation functions Ci are maximised by comparing the absolute values; a transmission path estimating part (140) for obtaining estimated values E|h = {Ehj} (j = 0, 1, ... L,) of impulse responses of the transmission path according to an adaptive equalising algorithm with respect to the sample value sequences and the corresponding training sequences thereto and from estimated values of the transmission symbol sequences and the sample value sequences ; and a Viterbi algorithm processing part (130) for obtaining the estimated values = {Exn} (n = 1, 2, ..., N) of transmission symbol sequences from the sample value sequences based on the estimated values E|h, characterised by sample phase smoothing means (104, 201, 301, 401) for obtaining a sampling phase τs based on the selected sampling phase obtained by said selecting means such that the bit error rate is minimised, a sampling processing part (120) for sampling the reception signals y(t) entered at specified timings with respect to a symbol interval T and the sampling phase τs to obtain sample value sequences = {yn} (n = 1, 2, ..., N). A maximum likelihood sequence estimator as set forth in claim 1, wherein: said selecting means (103) is configured to obtain an index Ik from the selected sampling phase designated τ(Ik), and said sampling phase smoothing means (104) is arranged to obtain a weighted means value, a maximum value and a minimum value of the index Ik, and to obtain an index Is by subtracting the minimum value from the maximum value to obtain a first resultant value, multiplying the first resultant value by a constant to obtain a second resultant value, and adding the second resultant value to the weighted mean value, the sampling phase τ s corresponding to the index Is. A maximum likelihood sequence estimator as set forth in claim 2, wherein said sampling phase smoothing means (104) is operable to obtain a weighted mean value, a maximum value and a minimum value of the index Ik every predetermined section. A maximum likelihood sequence estimator as set forth in claim 2 or 3, wherein: said sampling means (101) is configured to sample the reception signals y(t) at a symbol interval T with respect to respective sampling phases of simple non-increasing or simple non-decreasing sampling phase sequences τ = {τ(i)} (i = 1, 2, ..., M) to obtain the sample value sequences i, and said correlation value calculating means (102) is operable to obtain the complex cross-correlation functions Ci of the training sequences according to the following equation (where, i = 1, 2, ..., M: * means that complex conjunction is used). A maximum likelihood sequence estimator as set fort in claim 4 wherein: said sampling phase smoothing means (104) is arranged to obtain (1) the weighted means value, maximum value and minimum value of the index Ik for every predetermined section respectively as a sectional weighted mean value I avg (k), a sectional maximum value I max (k) and a sectional minimum value I min (k) of the index Ik, calculated according to the following equations at the time of the kth sampling phase selection: (where, wa (j) (j = 1, 2, ..., Ls) is a weighted coefficient and Ls denotes the maximum length of calculation), (2) a selective sampling phase index Is according to the following equation: Is = [I avg (k) + ws (I max (k) - I min (k))] (where, [ ] indicates the Gaussian notation which returns the largest integer not exceeding a value in the notation and ws denotes a shift coefficient), and (3) the sampling phase τs corresponding to the index Is according to the following equation: A maximum likelihood sequence estimator as set forth in claim 4 wherein: said sampling phase smoothing means (104) is arranged to obtain (1) the weighted mean value, maximum value and minimum value of the index Ik for every predetermined section respectively as a weighted means value I AVG (k), a maximum value I MAX (k) and a minimum value I MIN (k) of the index Ik, calculated according to the following equations at the time of the kth sampling phase selection: I AVG(k) = (1 - α) I AVG (k - 1) + αIk (α: constant) I AVG(O) = 0 (2) a selective sampling phase index Is according to the following equation: Is = [I AVG (k) + ws (I MAX (k) - I MIN (k))] (where, [ ] indicates the Gaussian notation which returns the largest integer not exceeding a value in the notation and ws denotes a shift coefficient), and (3) the sampling phase τs corresponding to the index Is according to the following equation: A maximum likelihood sequence estimator as set forth in claim 5 or 6, further comprising a delay processing part (110) for delaying the reception signals entered from the transmission lines as long a specified and outputting the signals to said sampling processing part, wherein: said sampling processing part (120) is operable to sample the reception signals from said delay processing part, said transmission estimating part (140) is arranged to obtain first estimated values E|h = {Ehj} (j = ), 1, ..., L) of the impulse responses of the transmission lines from the sample value sequences and training sequences corresponding to the sample value sequences , according to an adaptive equalisation algorithm, said transmission estimating part (140) is arranged to obtain second estimated values E h from the sample value sequences and the estimated values = {Exn} (n = 1, 2, ..., N) of the transmission symbol sequences according to the adaptive equalisation algorithm, and said operation processing part (130) is operable to obtain the estimated values according to a Viterbi algorithm. A maximum likelihood sequence estimator as set forth in claim 1, wherein said sampling phase smoothing means (301) is arranged to obtain a sectional weighted mean value, a sectional maximum value and a sectional minimum value, all of a sampling phase τsel(k) obtained as the selected sampling phase by said selecting means, and to obtain the sampling phase τs based on the sectional weighted mean value, the sectional maximum value and the sectional minimum value. A maximum likelihood sequence estimator as set forth in claim 8, wherein: said sampling means (101) is arranged to sample the reception signals at the symbol interval T, starting from an estimated leading time determined by a frame synchronisation mechanism, said sampling phase smoothing means is arranged to obtain a coordinate transformation term in accordance with the estimated leading time, add the coordinate transformation term to the sampling phase τsel(k), obtain the sectional weighted mean value, a sectional maximum value and a sectional minimum value of the sampling phase τsel(k) after adding the coordinate transformation term to the sampling phase τsel(k), and to calculate the sampling phase τs by respectively subtracting from and adding to the sectional weighted mean value, the coordinate transformation term and a second resultant value, the second resultant value being obtained by (1) subtracting the absolute value of the coordinate transformation term, and the sectional minimum value, from the sectional maximum value to obtain a first resultant value, and (2) multiplying the first resultant value by a constant to obtain the second resultant value. A maximum likelihood sequence estimator as set forth in claim 9, wherein: said sampling means (10) is arranged to sample the reception signals at the symbol interval T with respect to the respective sampling phases of the sampling phase sequences τ = {τ (i)} (i = 1, 2, ..., M), starting from an estimated leading time Et slot (k) determined by a frame synchronisation mechanism when processing the kth assigned time slot to obtain sample value sequences i = {yin} (i = 1, 2, ..., M; n = 1, 2, ..., Nt) of the reception signals, said correlation value calculating means (102) is configured to obtain the complex cross-correlation functions Ci of the training sequences based on the sample value sequences i and the training sequences = {n} (n = 1, 2, ..., Nt) of the sample value sequences yi, according to the following equation: (where, i = 1, 2, ..., M; * means that complex conjugation is used), and said sampling phase smoothing means is arranged to obtain (a) the coordinate transformation term as a coordinate transformation term tc(k), according to the equations given below: Δtc(k) = Et slot (k) - Et slot (k-1) k > 1 (b) the sectional weighted mean value, the sectional maximum value and the sectional minimum value respectively as a sectional weighted mean value τ avg (k), a sectional maximum value τ max (k) and a sectional minimum value τ min (k) of the sampling phase τsel(k), according to the following equations: (where, wa (j) (j = 1, 2, ..., Ls) is a weighted coefficient and Ls denotes the maximum calculation length), and (c) the selective sampling phase τs according to the following equation: τs = τ avg(k) - tc(k) + ws (τmax(k)-τmin(k) - |tc (k)|) (where, ws indicates the shift coefficient, wa (j) (j = 1, 2, ..., Ls) is a weighted coefficient and Ls denotes the maximum calculation length). A maximum likelihood sequence estimator as set forth in claim 8, wherein: said sampling means (101) is arranged to sample the reception signals at the symbol interval T, starting from an estimated leading time determined by a frame synchronisation mechanism, and said sampling phase smoothing means (301) is arranged to obtain first and second coordinate transformation terms in accordance with the estimated leading time, to add the second coordinate transformation term to the sampling phase τsel(k), to obtain the sectional weighted mean value, sectional maximum value and sectional minimum value of the sampling phase τsel(k) after adding the second coordinate transformation term to the sample phase τsel(k), and to calculate the sampling phase τsel by respectively subtracting the sectional minimum value and the absolute value of the first coordinate transformation term from the sectional maximum value to obtain a first resultant value, multiplying the first resultant value by a constant to obtain a second resultant value, and to add the second resultant value to the sectional weighted mean value. A maximum likelihood sequence estimator as set fort in claim 11, wherein: said sampling means (101) is arranged to sample the reception signals, at the symbol interval T with respect to the respective sampling phases of the sampling phase sequences τ = {τ (i)} (i = 1, 2, ..., M), starting from the estimated leading time slot Et slot (k) determined by the frame synchronisation mechanism when processing the kth assigned time slot to obtain the sample value sequences i = {yin} (i = 1, 2, ..., M; n = 1, 2, ..., Nt) of the reception signals y(t), said correlation value calculating means (102) is configured to obtain complex cross-correlation function Ci of the training sequences based on the sample value sequences i and the training sequences = {n} (n = 1, 2, ..., Nt) of the sample value sequences i, according to the following equation: (where, i = 1, 2, ..., M; * means that complex conjunction is used), and said sampling phase smoothing means (401) is configured to obtain (a) the first and second coordinate transformation terms, respectively as a first coordinate transformation term tc1(k) and a second coordinate transformation term tc2(j, k), according to the following equations at the time of the kth sample phase selection: (b) the sectional weighted means value, the sectional maximum value and the selectional minimum value, respectively as a sectional weighted mean value τ AVG (k), a sectional maximum value τ MAX (k) and a sectional minimum value τ MIN (k) of the sampling phase τsel(k), according to the following equations: (where, wa (j) (j = 1, 2, ..., Ls) is a weighted coefficient and Ls denotes the maximum calculation length), and (c) the selecting sampling phase τs according to the following equation: τs = τ AVG (k) + ws (τ MAX (k) - τ MIN (k) - |tc1 (k)|) (where, ws indicates the shift coefficient, wa (j) (j = 1,2, ..., Ls) is a weighted coefficient and Ls denotes the maximum calculation length). A maximum likelihood sequence estimator as set forth in claim 10, 11 or 12 further comprising a delay processing part (110) for delaying the reception signals and outputting the signals to said sampling processing part, wherein: the sampling phase sequences τ = {τ (i)} (i = 1, 2, ..., M) in said sampling means are simple non-increasing or simple non-decreasing, said sampling processing part (120) is operable to sample the reception signals from said delay processing part, said transmission estimating part (140) is arranged to obtain first estimated values E|h = {Ehj} (j = 0, 1, ..., L) of the impulse responses of the transmission path from the sample value sequences , according to an adaptive equalisation algorithm, said transmission estimating part (140) is arranged to obtain estimated values Eh from the sample value sequences and estimated values = {Exn} (n = 1, 2, ..., N) of transmission symbol sequences according to the adaptive equalisation algorithm, and said operation processing part (130) is operable to the estimated values according to a Viterbi algorithm. A maximum likelihood sequence estimating method comprising: sampling reception signals y(t) from a transmission path, at a symbol interval T with respect to respective sampling phases of sampling phase sequences τ = {T (i)} (i = 1, 2, ..., M), to obtain sample value sequences i = {yin} (i - 1, 2, ..., M; n = 1, 2, ..., Nt) of the reception signals y(t) obtaining complex cross-correlation functions = {Ci} (i = 1, 2, ..., M) of the sample value sequences i and training sequences = {n} (n = 1, 2, ..., Nt) of the sample value sequences i, and selecting a sampling phase at which the absolute values of the number M of complex cross-correlation functions Ci are maximised by comparing the absolute values, obtaining estimated values E|h = {Ehj} (j = 0, 1, ... L), of impulse responses of the transmission path according to an adaptive equalising algorithm with respect to the sample value sequences and the corresponding training sequence thereto and from estimated values of the transmission symbol sequences and the sample value sequences , and using the Viterbi algorithm to obtain the estimated values = {Exn} (n = 1, 2, ..., N) of transmission symbol sequences from the sample value sequences based on the estimated values E|h, characterised by obtaining a sampling phase τs based on the selected sampling phase such that the bit error rate is minimised, sampling the reception signals y(t) entered at specified timings with respect to a symbol interval T and the sampling phase τs to obtain sample value sequences = {yn} (n = 1, 2, ..., N). A method as set forth in claim 14, including obtaining an index Ik from the selected sampling phase designated τ(Ik), obtaining a weighted means value, a maximum value and a minimum value of the index Ik, obtaining an index Is by subtracting the minimum value from the maximum value to obtain a first resultant value, multiplying the first resultant value by a constant to obtain a second resultant value, and adding the second resultant value to the weighted mean value, so as to obtain the sampling phase τs corresponding to the index Is. A method as set forth in claim 15, including obtaining a weighted mean value, a maximum value and a minimum value of the index Ik every predetermined section. A method as set forth in claim 14, including sampling the reception signals y(t) at a symbol interval T with respect to respective sampling phases of simple non-increasing or simple non-decreasing sampling phase sequences τ = {τ(i)} (i = 1, 2, ..., M) to obtain the sample value sequences i, and obtaining the complex cross-correlation functions Ci of the training sequences , according to the following equation (where, i = 1, 2, ..., M: * means that complex conjunction is used). A method according to claim 17 including obtaining: (1) the weighted means value, maximum value and minimum value of the index Ik for every predetermined section respectively as a sectional weighted mean value I avg (k), a sectional maximum value I max (k) and a sectional minimum value I min (k) of the index Ik, calculated according to the following equations at the time of the kth sampling phase selection: (where, wa (j) (j = 1, 2, ..., Ls) is a weighted coefficient and Ls denotes the maximum length of calculation), (2) a selective sampling phase index Is according to the following equation: Is = [I avg (k) + ws (I max (k) - I min (k))] (where, [ ] indicates the Gaussian notation which returns the largest integer not exceeding a value in the notation and ws denotes a shift coefficient), and (3) the sampling phase τs corresponding to the index Is according to the following equation: A method as set forth in claim 16 including obtaining: (1) the weighted mean value, maximum value and minimum value of the index Ik for every predetermined section respectively as a weighted means value I AVG (k), a maximum value I MAX (k) and a minimum value I MIN (k) of the index Ik, calculated according to the following equations at the time of the kth sampling phase selection: I AVG(k) = (1 - α) I AVG (k - 1) + αIk (α: constant) I AVG(O) = 0 (2) a selective sampling phase index Is according to the following equation: Is = [I AVG (k) + ws (I MAX (k) - I MIN (k))] (where, [ ] indicates the Gaussian notation which returns the largest integer not exceeding a value in the notation and ws denotes a shift coefficient), and (3) the sampling phase τs corresponding to the index Is according to the following equation: A method as set forth in claim 19, including obtaining a sectional weighted mean value, a sectional maximum value and a sectional minimum value, all of a sampling phase τsel(k) obtained as the selected sampling phase by said selecting means, and to obtain the sampling phase τs based on the sectional weighted mean value, the sectional maximum value and the sectional minimum value. A method as set forth in claim 20, including: sampling the reception signals at the symbol interval T, starting from an estimated leading time determined by a frame synchronisation mechanism, obtaining a coordinate transformation term in accordance with the estimated leading time, adding the coordinate transformation term to the sampling phase τsel(k), obtaining the sectional weighted mean value, a sectional maximum value and a sectional minimum value of the sampling phase τsel(k) after adding the coordinate transformation term to the sampling phase τsel(k), and to calculating the sampling phase τs by respectively subtracting from and adding to the sectional weighted mean value, the coordinate transformation term and a second resultant value, the second resultant value being obtained by (1) subtracting the absolute value of the coordinate transformation term, and the sectional minimum value, from the sectional maximum value to obtain a first resultant value, and (2) multiplying the first resultant value by a constant to obtain the second resultant value. A method as set forth in claim 21, wherein: sampling the reception signals at the symbol interval T with respect to the respective sampling phases of the sampling phase sequences τ = {τ (i)} (i = 1, 2, ..., M), starting from an estimated leading time Et slot (k) determined by a frame synchronisation mechanism when processing the kth assigned time slot to obtain sample value sequences i = {yin) (i = 1, 2, ..., M; n = 1, 2, ..., Nt) of the reception signals, obtaining the complex cross-correlation functions Ci of the training sequences based on the sample value sequences yi and the training sequences = {n} (n = 1, 2, ..., Nt) of the sample value sequences i, according to the following equation: (where, i = 1, 2, ..., M; * means that complex conjugation is used), and obtaining (a) the coordinate transformation term as a coordinate transformation term tc(k), according to the equations given below: Δtc(k) = Et slot (k) - Et slot (k-1) k > 1 (b) the sectional weighted mean value, the sectional maximum value and the sectional minimum value respectively as a sectional weighted mean value τ avg (k), a sectional maximum value τ max (k) and a sectional minimum value τ min (k) of the sampling phase τsel(k), according to the following equations: (where, wa (j) (j = 1, 2, ..., Ls) is a weighted coefficient and Ls denotes the maximum calculation length), and (c) the selective sampling phase τs according to the following equation: τs = τ avg(k) - tc(k) + ws(τmax(k)-τmin(k) - |tc (k)|) (where, ws indicates the shift coefficient, wa (j) (j = 1, 2, ..., Ls) is a weighted coefficient and Ls denotes the maximum calculation length). A method as set forth in claim 20, including: sampling the reception signals at the symbol interval T, starting from an estimated leading time determined by a frame synchronisation mechanism, and obtaining first and second coordinate transformation terms in accordance with the estimated leading time, adding the second coordinate transformation term to the sampling phase τsel(k), obtaining the sectional weighted mean value, sectional maximum value and sectional minimum value of the sampling phase τsel(k) after adding the second coordinate transformation term to the sample phase τsel(k), and calculating the sampling phase τsel by respectively subtracting the sectional minimum value and the absolute value of the first coordinate transformation term from the sectional maximum value to obtain a first resultant value, multiplying the first resultant value by a constant to obtain a second resultant value, and to add the second resultant value to the sectional weighted mean value. A maximum likelihood sequence estimator as set forth in claim 22, including: sampling the reception signals entered from the transmission lines, at the symbol interval T with respect to the respective sampling phases of the sampling phase sequences τ = {τ (i)} (i = 1, 2, ..., M), starting from the estimated leading time slot Et slot (k) determined by the frame synchronisation mechanism when processing the kth assigned time slot to obtain the sample value sequences yi = {yin} (i = 1, 2, ..., M; n = 1, 2, ..., Nt) of the reception signals y(t), obtaining complex cross-correlation function Ci of the training sequences based on the sample value sequences yi and the training sequences = {n} (n = 1, 2, ..., Nt) of the sample value sequences i, according to the following equation: (where, i = 1, 2, ..., M; * means that complex conjunction is used), and obtaining (a) the first and second coordinate transformation terms, respectively as a first coordinate transformation term tc1(k) and a second coordinate transformation term tc2(j, k), according to the following equations at the time of the kth sample phase selection: (b) the sectional weighted means value, the sectional maximum value and the selectional minimum value, respectively as a sectional weighted mean value τ AVG (k), a sectional maximum value τ MAX (k) and a sectional minimum value τ MIN (k) of the sampling phase τsel(k), according to the following equations: (where, wa (j) (j = 1, 2, ..., Ls) is a weighted coefficient and Ls denotes the maximum calculation length), and (c) the selecting sampling phase τs according to the following equation: τs = τ AVG (k) + ws (τ MAX (k) - τ MIN (k) - |tc1 (k)|) (where, ws indicates the shift coefficient, wa (j) (j = 1,2, ..., Ls) is a weighted coefficient and Ls denotes the maximum calculation length). A method as set forth in claim 18, 19, 21 or 23 including: delaying the reception signals, sampling the delayed reception signals, obtaining first estimated values E|h = {Ehj} (j = 0, 1, ..., L) of the impulse responses of the transmission path from the sample value sequences and training sequences corresponding to the sample value sequences , according to an adaptive equalisation algorithm, obtaining estimated values Eh from the sample value sequences and estimated values = {Exn} (n = 1, 2, ..., N) of transmission symbol sequences according to the adaptive equalisation algorithm, and obtaining the estimated values according to a Viterbi algorithm.

Inventors: SHIINO HARUHIRO, YAMAGUCHI NORIO, SHIINO, HARUHIRO, YAMAGUCHI, NORIO, SHIINO, HARUHIRO OKI ELECTRIC INDUSTRY CO., LTD., YAMAGUCHI, NORIO OKI ELECTRIC INDUSTRY CO., LTD.
Applicants: OKI ELECTRIC IND CO LTD, OKI ELECTRIC INDUSTRY COMPANY, LIMITED
Application year: 1993
Main classification: H04L7
IPC codes: H03M13, H04L7, H04L25, H04B3