Bias circuit for a memory line decoder driver of non-volatile memories

Abstract: A memory line decoding driver (1) is so biased that the P channel pull-up transistor (6) biasing the final inverter (5) conducts a high current during the line address transient phase, for rapidly charging the input of the final inverter, and is turned on weakly during the static phase between one address phase and another, for reducing current consumption. For which purpose, a voltage modulating stage (18) alternatively connects the gate terminal of the pull-up transistor (6) to a capacitor (37), with which the charge is distributed, and to the supply (VPC).

Description: The present invention relates to a memory line decoder driver for nonvolatile memories, memory line here being intended to mean a row or column of the memory. As is known, row and/or column decoder drivers provide for driving associated rows or columns on the basis of a selection signal and address signals by which the associated row or column address is coded. Like other nonvolatile memory components, drivers of the aforementioned type present conflicting requirements of low consumption and high reading speed. In fact, the speed at which the rows and columns in the memory array are addressed (and hence the speed of the overall reading phase) increases alongside an increase in the current charging the row or column capacitance, thus resulting in increased consumption. Though the following description applies to both row and column decoder drivers, for the sake of simplicity reference will be made herein to a row decoder driver. To enable a clearer understanding of the problem, a description will first be given of a typical row driver arrangement with reference to Figure 1 in which the driver is indicated as a whole by 1. As shown in Figure 1, driver 1 comprises a NAND circuit 2 with three inputs for coding the address of the row to be addressed (signals A, B and C); a selection transistor 3 interposed between the output of NAND circuit 2 and a node 4, and the gate terminal of which is supplied with a selection signal Pi; a final inverter 5 with its input connected to node 4 and its output 7 connected to the word line WL associated with the driven row; and a P channel pull-up transistor 6 interposed between a supply line VPC and node 4, and the gate terminal of which is supplied with a voltage control signal V1. In the Figure 1 circuit, pull-up transistor 6 is required for maintaining the input of final inverter 5 biased when selection signal Pi is maintained low, i.e. for deselecting the group of rows addressed by Pi (the same selection signal Pi is also supplied to other rows, generally sixteen in all). In this case, in fact, signal Pi is switched to low, so that selection transistor 3 is disabled, and, in the absence of a pull-up transistor, would leave the input of final inverter 5 floating. Pull-up transistor 6 on the other hand provides for connecting node 5 to supply line VPC and so correctly biasing the input of final inverter 5. At present, pull-up transistor 6 is driven in two ways. In a first known solution, signal V1 is such as to keep it on permanently, whereas, in a second known solution, signal V1 switches according to whether the associated row is selected or deselected, and more specifically turns off pull-up transistor 6 if the associated row is selected, as shown more clearly in Figure 2. In Figure 2 (in which the same parts as in Figure 1 are indicated using the same numbering system), the gate terminal of pull-up transistor 6 is connected to the output of an inverter 10, the input of which is connected to input node 4 of final inverter 5. Both the solutions described above present drawbacks. In fact, both present zero consumption under static conditions (at the end of the address switching phase -switch transient-) in the event the associated row is not selected, but behave differently at the end of the row selection transient (a row is selected when inputs A-C and selection signal Pi of its driver are all high). The first solution in fact (wherein pull-up transistor 6 is always on) presents a current path between pull-up transistor 6 (on), selection transistor 3 and the N channel transistors (not shown) forming NAND circuit 2; which current path, in view of the fairly large size of the driver components for ensuring fast operation, results in far from negligible consumption. In the second solution (Figure 2 circuit) on the other hand, following the selection transient (node 4 low), pull-up transistor 6 is turned off so that consumption is zero under static conditions. The second solution, however, requires the use of a carefully designed feedback circuit which may present operating problems in the event of a wide variation in the supply voltage of driver 1; and, what is more, involves a fairly complex, cumbersome layout. It is an object of the present invention to provide a driver bias circuit designed to overcome the drawbacks of known solutions, and which in particular is such as to provide for low consumption of the driver under static conditions, and high-speed switching during the selection transient. According to the present invention, there is provided a bias circuit for the memory line decoder driver of nonvolatile memories, as claimed in Claim 1. In practice, according to the present invention, the bias voltage of the pull-up transistor is so modulated as to be reduced during address selection, so as to achieve high conduction of the pull-up transistor and rapidly charge the inputs of the final inverters of the rows to be deselected; and is maintained at a higher level under static conditions, so as to keep the pull-up transistor turned on weakly and so minimize consumption of the driver of the selected rows. A preferred, non-limiting embodiment of the present invention will be described by way of example with reference to the accompanying drawings, in which: Figure 1 shows a known type of row decoder driver; Figure 2 shows the same driver featuring a known bias circuit; Figure 3 shows the bias circuit according to the present invention; Figure 4 shows a graph of two quantities in Figure 3. In Figure 3, the bias circuit is indicated as a whole by 15, and substantially comprises a monostable stage 16, a level shift device 17, and a voltage modulating stage 18. More specifically, monostable stage 16 presents two inputs 20 and 21 supplied respectively with signals ATD and ENN. Signal ATD is a logic signal (see also Figure 4) which switches to high upon switching of the first row address in the memory array (not shown) of which bias circuit 15 and driver 1 form part, and switches back to low upon switching of the last row address. Signal ENN is an inverted enabling signal which only permits operation of bias circuit 15 when low. Inputs 20 and 21 are connected to a NOR circuit 22, the output of which is connected to the input of an inverter 23 and to a capacitor 24 interposed between the output of the NOR circuit and ground. The output of inverter 23 is connected to one input of a further NOR circuit 25, the second input of which is connected to input 20 of the circuit. The output of NOR circuit 25 (presenting logic signal IN) is connected to input 27 of level shift device 17, which provides for interfacing the logic circuits supplied with voltage V DD (not shown, supplying monostable stage 16) and those of the row driver which are supplied with voltage VPC. Level shift device 17 presents two outputs 28 and 29 respectively presenting an OUT signal and an OUTN signal inverted in relation to OUT. More specifically, the OUT signal is at 0 V (and the OUTN signal at VPC) when the IN signal at input 27 is low; and is at VPC (and the OUTN signal at 0 V) when the IN signal is high, according to the following table: IN OUT OUTN 0 0 V VPC 1 VPC OV Voltage modulating stage 18 comprises five transistors M1-M5, of which M1 is a P channel transistor, and M2-M5 N channel transistors. More specifically, transistor M1 presents a source terminal connected to supply line VPC; a gate terminal defining input 33 of circuit 15 and supplied with signal DE (for disabling the associated pull-up transistor 6); and a drain terminal connected to the output node 34 of circuit 15 supplying the bias voltage V1 of pull-up transistor 6 of circuit 1 in Figure 1. Transistor M2 presents a drain terminal connected to supply line VPC; a gate terminal connected to output 28 of level shift device 17; and a source terminal connected to node 34. Transistor M3 presents a drain terminal connected to node 34; a gate terminal connected to output 29 of level shift device 17; and a source terminal connected to node 35. Transistor M4 presents a drain terminal connected to node 35; a gate terminal connected to a bias line 36 (e.g. 1.9 V); and a grounded source terminal. Transistor M5 presents a drain terminal connected to node 34; a gate terminal connected to bias line 36; and a grounded source terminal. Transistors M1-M5 are not all the same size. More specifically, M4 and M5 are substantially longer and hence far more resistive as compared with M2 and M3 for the reasons given below. Voltage modulating stage 18 also comprises a capacitor 37 interposed between node 35 and ground. The Figure 3 circuit operates as follows (refer also to Figure 4). Transistors M4 and M5 are always on, and, during normal operation of the circuit, DE is high (so that M1 is off) and ENN is low. Under static conditions, signal ATD is also low, so that the output of NOR circuit 22 is high, the output of inverter 23 is low, IN is high, the OUT signal is at VPC, the OUTN signal is at 0 V, and consequently M2 is on and M3 off. In the above condition, capacitor 37 is short-circuited to ground by M4, and M2, M5 operate as a resistive divider wherein, in view of the difference in size and hence equivalent resistance of the two transistors, node 34 presents a voltage V1 H according to the equation: V1 H = VPC - VT M2 where VT M2 is the threshold voltage of transistor M2 (see also Figure 4). In the above condition, the pull-up transistors 6 connected to bias circuit 15 are turned on weakly, thus minimizing (to about 10µA) the current consumption of pull-up transistors 6 of the selected rows under static conditions. As regards the deselected rows, pull-up transistors 6 are turned on, albeit weakly, by a sufficient amount to maintain a high bias state at the input of respective final inverters 5 with no consumption on the part of the pull-up transistors. When address switching, signal ATD presents a pulse (Figure 4) which activates monostable stage 16. More specifically, upon signal ATD switching to high, NOR circuit 25 switches, the IN signal switches to low, level shift device 17 switches, the OUT signal switches to low, the OUTN signal switches to high (VPC), so that M2 is turned off and M3 turned on. As node 34 is no longer related to supply line VPC, the charge is distributed between the gate regions of pull-up transistors 6 connected to node 34 and capacitance C1 of capacitor 37 (initially discharged). The circuit in fact is in a condition in which the capacitances of the pull-up transistor gate regions are in series with capacitor 37, via M3, thus forming a capacitance divider (in this phase, M4 and M5 are practically ineffective by virtue of their high equivalent resistance). As a result, voltage V1 at node 34 falls to a predetermined level according to the equation: V1 L = VPC * C pu /(C pu + C1) where C pu is the capacitance of the gate region of pull-up 6, thus enabling increased current conduction by all the pull-up transistors connected to node 34, for rapidly charging the inputs of final inverters 5. The monostable stage switches shortly after zero switching of ATD. In fact, when ATD switches, capacitor 24 begins charging, at the end of which signal IN switches so as to turn off transistor M3, turn on transistor M2, and so restore the static conditions wherein V1 is restored to V1 H , and capacitor 37 discharges groundwards via M4. The delay in the switching of monostable stage 16 in relation to signal ATD provides for ensuring all the inputs of the final inverters of the deselected rows are charged correctly. Signal DE, which is normally high, provides, when switched to low, for simultaneously turning off all the pull-up transistors connected to node 34 (e.g. when all the word lines in the sector or memory array are addressed, and so eliminating consumption due to conduction of the pull-up transistors which, albeit low, would be significant over a large number of transistors). When high, the enabling signal provides for grounding voltage V1, in which case, the IN output of monostable stage 16 is low, outputs OUT and OUTN are respectively low and high, transistor M2 is turned off and M3 turned on, thus resulting in a situation similar to that of the selection transient, except that, in this case, the phase is much longer and hence the effect more noticeable of M4 and M5 which discharge capacitor 37 and the gate capacitance of the pull-up transistors until V1 is zeroed. The advantages of the circuit according to the present invention will be clear from the foregoing description. Firstly, it provides for optimum performance in terms of consumption and speed. In fact, static consumption for the selected rows is considerably reduced by virtue of the pull-up transistors being turned on only weakly, for which reason it is also possible to make the pull-up transistors smaller as compared with known drivers. In addition to a reduction in area, this therefore provides for a further reduction in consumption, and with no falloff in speed, in that, by modulating the bias voltage, the pull-up transistors are still capable, during the deselection transients, to conduct a strong current for rapidly charging the inputs of the inverters. Secondly, the solution described is straightforward, and may be implemented easily, as a separate circuit for biasing a large number of drivers, with no alteration to the layout of the drivers as required formerly for the Figure 2 circuit. On the whole, therefore, the layout of the drivers and relative bias circuit is greatly simplified. Finally, by virtue of the solution described being independent of the row decoder supply voltage, it may also be applied to devices operating within a fairly wide supply voltage range. To those skilled in the art it will be clear that changes may be made to the circuit as described and illustrated herein without, however, departing from the scope of the present invention. In particular, it may be applied equally as well to column drivers; and the monostable stage and voltage modulator configurations may vary as compared with those illustrated.

Claims: A bias circuit (15) for a memory line decoder driver (1) of nonvolatile memories, wherein said driver (1) comprises final drive means (5) , and pull-up transistor means (6) connected between a first reference potential line (VPC) and the input (4) of said final drive means; said drive means (5) being designed to switch during address transition phases, and to maintain a steady state during static phases interposed between said address transition phases; said bias circuit (15) comprising generating means (18) for generating a variable bias signal (V1) for said pull-up transistor means (6); characterized by the fact that said generating means (18) comprise switching means (M1-M5) for switching said bias signal (V1) between a first level (V 1H ) corresponding to a weak turn-on state of said transistor means (6) at least during said static phases, and a second level (V 1L ) corresponding to a high turn-on state of said transistor means (6) during said address transition phases. A bias circuit as claimed in Claim 1, characterized by the fact that it comprises a monostable stage (16) having an input (20) supplied with an address transition detection signal (ATD), and designed to generate a switch signal (IN) for said generating means (18). A bias circuit as claimed in Claim 2, characterized by the fact that said monostable stage (16) comprises means (22-25) for generating said switch signal (IN) activated by, and of longer duration than, said address transition detection signal (ATD). A bias circuit as claimed in Claim 3, characterized by the fact that said monostable stage (16) comprises logic circuits. A bias circuit as claimed in Claim 3 or 4, characterized by the fact that said monostable stage (16) presents a second input (21) supplied with an enabling signal (ENN) for switching said monostable stage and said bias circuit (15) between an enabled state wherein said bias signal (V1) switches between said first and second level, and a disabled state wherein said bias signal corresponds to a full-on state of said transistor means (6). A bias circuit as claimed in one of the foregoing Claims from 2 to 5, characterized by the fact that it comprises level shift means (17) interposed between said monostable stage (16) and said generating means (18). A bias circuit as claimed in one of the foregoing Claims from 2 to 6, characterized by the fact that said switching means (M1-M5) comprise first switch means (M2) interposed between said first reference potential line (VPC) and an output node (34); second switch means (M3) interposed between said output node and a signal level switch node (35), said first and second switch means (M2, M3) being push-pull controlled by said switch signal (IN); capacitive means (37) interposed between said signal level switch node (35) and a second reference potential line; and discharge means (M4), interposed between said signal level switch node (35) and said second reference potential line, for connecting said output node (34) to a potential close to said first reference potential line during said static phases, and for connecting said output node (34) to said capacitive means during said transient phases. A circuit as claimed in Claim 7, characterized by the fact that said first and second switch means (M2, M3) and said discharge means (M4) respectively comprise a first, second and third MOS transistor; said third MOS transistor being much longer than said first and second MOS transistors. A circuit as claimed in Claim 8, characterized by the fact that said switching means also comprise a fourth MOS transistor (M5) interposed between said output node (34) and said second reference potential line; said fourth MOS transistor being much longer than said first and second MOS transistors (M2, M3); said third and fourth MOS transistors (M4, M5) having gate terminals connected to a third reference potential line (36). A circuit as claimed in one of the foregoing Claims from 7 to 9, characterized by the fact that said switching means also comprise a fifth MOS transistor (M1) interposed between said first reference potential line (VPC) and said output node (34). A circuit as claimed in Claims 9 and 10, characterized by the fact that said first, second, third and fourth MOS transistors (M2-M5) are N channel transistors, and said fifth transistor (M1) is a P channel transistor. A memory line decoding driver (1) for nonvolatile memories, comprising final drive means (5), and pull-up transistor means (6) connected between a first reference potential line (VPC) and the input (4) of said final drive means; said final drive means (5) being designed to switch during transient selection and deselection phases, and to maintain a steady state during static phases interposed between said transient phases; characterized by the fact that it comprises a bias circuit (15) as claimed in any one of the foregoing Claims from 1 to 11. A method of biasing the memory line decoding driver (1) of nonvolatile memories, said driver comprising final drive means (5), and pull-up transistor means (6) connected between a first reference potential line (VPC) and the input (4) of said final drive means; said final drive means (5) being designed to switch during address transition phases, and to maintain a steady state during static phases interposed between said address transition phases; said method comprising the step of generating a variable bias signal (V1) for said pull-up transistor means; characterized by the fact that said generating step comprises the step of switching said bias signal (V1) between a first level (V 1H ) corresponding to a weak turn-on state of said transistor means (6) at least during said static phases, and a second level corresponding to a high turn-on state of said transistor means during said address transition phases.

Inventors: GOLLA CARLA MARIA, MACCARRONE MARCO, PASCUCCI LUIGI, GOLLA,CARLA MARIA, MACCARRONE,MARCO, PASCUCCI,LUIGI
Applicants: SGS THOMSON MICROELECTRONICS, SGS-THOMSON MICROELECTRONICS S.R.L.
Application year: 1993
Main classification: G11C8
IPC codes: G11C17, G11C16, G11C8