Lamp ballast circuit

Abstract: A ballast circuit having a series inductor (L₇) and capacitor (C₁₀) in which the lamp load (LL) is connected in parallel with the capacitor. During pre-ignition of the lamp load, the driving signal supplied by an inventor generating a substantially rectangular signal includes a fundamental frequency f₁. The resonant frequency f₀ of the series connected L-C circuit is at least 2 times greater than the fundamental frequency f₁ but less than the third harmonic of the driving signal.

Description: BACKGROUND OF THE INVENTION This invention relates to a ballast circuit for generating a substantially rectangular driving signal sufficient to ignite a lamp load, comprising: inductor means; capacitor means serially connected to said inductor means; and generating means for applying a generated signal to said serially connected inductor means and capacitor means, said generated signal having at least a fundamental frequency f 1 ; the inductor means and capacitor means having a resonant frequency f o . Inductor means are to understand to be means adapted to exhibit the properties of an inductor. Capacitor means are to understand to be means adapted to exhibit the properties of a capacitor. Conventionally the lamp load is connected across the capacitor. In a known circuitry the series L-C circuit operates during pre-ignition of the lamp load substantially at its resonant frequency. That is, the driving signal applied to the series L-C circuit is at or near the resonant frequency of the series L-C circuit. In this way a sufficiently high pre-ignition voltage is applied across the lamp load for ignition of the latter. The lamp load, typically of a fluorescent type, following ignition, achieves a substantially steady-state sinusoidal current flow therethrough by reducing the driving signal frequency well below the resonant frequency of the series L-C circuit. In determining when to switch from the resonant frequency to a different steady-state operating frequency, feedback circuitry is required in the known ballast circuit for sensing lamp ignition. A sufficiently high voltage during pre-ignition of the lamp and sinusoidal lamp current following ignition (i.e. steady state operation), is commonly provided by a bridge inverter. Both full bridge and half-bridge inverters are known in the ballast circuit art. The (half)-bridge inverter includes switching to control the frequency of the driving signal applied to the series L-C circuit. Control circuitry, responsive to the feedback circuitry, is required for controlling the speed at which the switching takes place. Known lamp ballast circuits, as described above, suffer from several drawbacks. For example, known lamp ballast circuits require generating two different frequencies, that is, the resonant frequency during pre-ignition of the lamp load and a different therefrom a steady-state operating frequency. Such ballast circuits also require sensing circuitry to determine when to switch from the resonant frequency to the steady state operating frequency. It is particularly undesirable to operate at or near the resonant frequency of the series L-C circuit before lamp ignition inasmuch as unsafe, high voltages and current levels can occur (i.e. above the maximum ratings of one or more ballast circuit components). By operating below resonance during pre-ignition of the lamp load, capacitive switching of the inverter can easily occur producing high switching losses. Additional circuitry is therefore required to prevent the inverter from operating below the series L-C circuit resonant frequency during pre-ignition of the lamp load. The inductance of inductor L is normally determined based on the desired lamp current during steady state conditions. The capacitance of capacitor C is thereafter chosen so as to provide a resonant condition (typically between 20-50 kHz for a fluorescent lamp). Generally, the capacitance of capacitor C is between about 5 to 10 nanofarads with the additional high voltage capability leading to a relatively costly capacitor requiring a relatively large space on a printed circuit board. Accordingly, it is desirable to provide a lamp ballast circuit having a safe open circuit (i.e., pre-ignition) voltage and current level, with relatively low switching losses. The improved lamp ballast circuit should not need a driving signal at more than one frequency, this frequency being well below resonance of the series L-C circuit. It is also desirable that the improved lamp ballast circuit permit use of a relatively less expensive, smaller capacitor in order to lower the lamp ballast manufacturing cost and to reduce the reactive current flowing through the capacitor after lamp ignition thus lowering circuit power loss. SUMMARY OF THE INVENTION In accordance with the invention, a ballast circuit for generating a driving signal sufficient to ignite a lamp-load as mentioned in the preamble is characterized in that for the fundamental frequency f 1 and the resonant frequency f o it holds: nf 1 < f o < (n+1)f 1 with n = an even integer. By operating in these regions during pre-ignition, safe voltage and current levels will be maintained. A single drive frequency results in safe non-resonant operation before lamp ignition as well as correct lamp current after ignition. Feedback circuitry for sensing ignition of the lamp load for switching to a different steady-state lamp operating frequency need not be provided. By eliminating the need to operate at the resonant frequency of the series connected L-C circuit during pre-ignition of the lamp load, the value and resulting size of the capacitor can be chosen far smaller than normally used in a conventional series connected L-C circuit in a known ballast circuit. In accordance with a feature of the invention, the generated signal which is a train of square waves, is generated preferably by a half-bridge or full bridge inverter. In yet another feature of the invention, the resonant frequency of the series connected L-C circuit is less than the third harmonic frequency of the generated square wave drive thereby avoiding unsafe third harmonic voltages and current levels during pre-ignition of the lamp load. Substantially the same generated signal frequency is used during pre-ignition and steady-state operation of the lamp load. Accordingly, it is an object invention to provide an improved ballast circuit in which the unloaded, open circuit voltage and current levels are within the operating range of the ballast circuit components. It is another object of the invention to provide an improved ballast circuit in which the same inverter driving signal can be used during pre-ignition and steady-state operation of the lamp load. It is a further object of the invention to provide an improved ballast circuit in which less costly components can be used to lower the manufacturing cost of the ballast. It is still another object of the invention to provide an improved ballast circuit which eliminates the need for feedback circuitry for sensing lamp ignition for changing the inverter frequency. It is still a further object of the invention to provide an improved ballast circuit in which the inverter driving signal frequency is substantially less than the resonant frequency of a series connected L-C output circuit during pre-ignition of the lamp load. The invention accordingly comprises several steps in a relation of one or more of such steps with respect to each of the others, and the device embodying features of construction, a combination of elements and arrangement of parts which are adapted to effect such steps, all is exemplified in the following detailed disclosure and the scope of the invention will be indicated in the claims. BRIEF DESCRIPTION OF DRAWINGS For a fuller understanding of the invention, reference is made to the following description taken in connection with the accompanying drawings, in which: Fig. 1 is a circuit diagram of a ballast output circuit in accordance with the present invention; Figs. 2(a), 2(b) and 2(c) are timing diagrams of an inverter substantially rectangular output voltage, output current at its fundamental frequency and output current at its third harmonic, respectively in the circuit according to Fig. 1; Fig. 3 is a schematic diagram of a ballast circuit in accordance with the invention; Figs. 4(a), 4(b), 4(c) and 4(d) are timing diagrams of signals produced within the ballast circuit of Fig. 3 during pre-ignition and steady-state operation of the lamp load; and Fig. 5 is a diagram of simulation of current in circuit of Fig. 1 as function of the ratio fundamental frequency and resonant frequency. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT The figures shown herein illustrate a preferred embodiment of the invention. Those elements/components shown in more than one figure of the drawings have been identified by like reference numerals/letters and are of similar construction and operation. Referring now to Figs. 1, 2(a), 2(b) and 2(c), a ballast circuit having a ballast output circuit 10 includes an inductor L and a capacitor C serially connected across the output of a square wave generator 13. Square wave generator 13 is preferably, but not limited to, a bridge inverter generating a substantially square wave of voltage ±E (i.e. the inverter output voltage). A lamp load 16 is connected across capacitor C through a switch SW. A current I flowing through inductor L includes a fundamental frequency component I f1 and a third harmonic component of the fundamental frequency I 3f1 . Other currents at higher odd harmonics are present but are significantly smaller. For the sake of simplicity in calculations with respect to the preferred embodiment as described hereafter only terms concerning the fundamental frequency f 1 and the 3rd harmonic are taken into account. In accordance with the Fourier transform square wave voltage 13 contains a sinusoidal wave at a fundamental frequency f 1 and odd harmonics of the fundamental frequency including a sinusoidal wave at a third harmonic 3f 1 . The amplitude of third harmonic component f 1 of voltage E is one third the amplitude of fundamental frequency component f 1 of voltage E. To achieve low switching losses within square wave generator 13 during pre-ignition of lamp load 16 (generally at trailing edges E T of voltage E), current I is preferably inductive (i.e., current lagging drive voltage) rather than capacitive (i.e. current leading drive voltage) during the voltage transitions of voltage E. Accordingly, the sum of fundamental frequency current component I f1 and third harmonic-current component I 3f1 is inductive wherein I 1f and I 3f1 are the capacitive and inductive components of I, respectively. To achieve an overall inductive current I, an impedance Z of circuit 10 as viewed from square wave generator 13 requires that the inductive impedance at the third harmonic Z 3f1 be less than one third the capacitive impedance at the fundamental frequency Z f1 . In other words, third harmonic component current I 3f1 is greater than fundamental frequency component I f1 . This relationship is illustrated in Figs. 2(b) and 2(c) wherein an amplitude P represents the peak value of fundamental frequency current component I f1 but is less than the peak value of third harmonic current component I 3f1 . In this way the sum of I f1 and I 3f1 remains inductive at the voltage transitions of voltage E. Lamp load 16 prior to ignition (i.e. during pre-ignition) appears as an open circuit. This open circuit condition is represented by switch SW in an open state (turned OFF). Following ignition, lamp load 16 is in its steady-state mode of operation and is represented by switch SW being turned ON such that lamp load 16 is connected in parallel with capacitor C. Impedance Z 3f1 , which must be less than one third impedance Z f1 during pre-ignition of lamp load 16, is therefore based on switch SW in its open state (i.e., turned OFF). This condition can be expressed as follows: ¦Z f1 ¦ > ¦3 Z 3f1 ¦ That is, ¦2πf 1 xL-1/(2πf 1 xC)¦ > 3 ¦6πf 1 xL-1/(6πf 1 xC)¦ Since impedance Z is capacitive at fundamental frequency f 1 and inductive at the third harmonic 3f 1 , 1/(2πf 1 xC)-2πf 1 xL > 18πf 1 xL - 1/(2πf 1 xC) That is, 1/(2πf 1 xC) > 5(2πf 1 xL) Eq. 3 can be rewritten as follows: 1/√ LC > √5 2πf 1 A resonant frequency f 0 of circuit 10 during pre-ignition (i.e., with switch SW open) can be defined as follows: 1/ √LC = 2π f 0 Substituting the value of 1/√LC defined by eq. 4 for the value of 1/√LC in eq. 5 results in 2πf 0 > √5 2πf 1 Accordingly, resonant frequency f 0 can be expressed as follows: f 0 > √5 f 1 In other words, third harmonic inductive current component I 3f1 is greater than fundamental frequency capacitive current component I f1 when resonant frequency f 0 is greater than √5 times the fundamental frequency of voltage E. To ensure that unsafe voltages and currents present at resonant frequency f 0 cannot occur, resonant frequency f 0 also should be less than third harmonic frequency 3f 1 of voltage E. Therefore, the values of inductor L and capacitor C should be chosen such that: √5f 1 < f 0 < 3f 1 By designing ballast circuit 10 such that resonant frequency f 0 is within the range of frequencies defined by eq. 8, the unsafe voltages and currents which occur at resonant frequency f 0 during pre-ignition of lamp load 16 are avoided and total current delivered by square wave generator 13 remains inductive. There is no need to vary the frequency of voltage E between resonant frequency f 0 during pre-ignition of lamp load 16 and a different frequency immediately thereafter as in conventional ballast circuitry. Feedback circuitry designed to sense ignition of lamp load 16 for determining when to vary the frequency of voltage E from resonant frequency f 0 to a different operating frequency can be eliminated. In accordance with the invention, a safer, simpler circuit is provided by maintaining resonant frequency f 0 within the boundaries defined by eq. 8. Due to the fact that the calculation as shown has only taken into account the fundamental frequency f 1 and its 3rd harmonic 3 f1 , the lower value of the range for chosing the resonant frequency f 0 is √5 times f 1 . However, when taken into account the existence of higher harmonics this value reaches the limit 2. The result of a simulation in which at least the first 25 harmonics are taken into account is shown in figure 5. In Fig. 5 the depicted curve displays the total current I t=o in the circuit of Fig. 1 at the moment the voltage switches from -E to +E of the generator 13 as function of ratio of the fundamental frequency f 1 and the resonant frequency f o . The circuit operates in the inductive mode in all those regions that the current I t=o is lagging to the voltage, thus is negative. From the Fig. 5 it is clear that these regions fulfil the relation nf 1 < f o < (n+1)f 1 with n = an even integer. A ballast circuit 20 in accordance with the invention is shown in Fig. 3. An input voltage of 277 volts, 60 hertz is supplied to an electromagnetic interference (EMI) suppression filter 23. Filter 23 filters high frequency components inputted thereto lowering conducted and radiated EMI. The output of filter 20 provided at a pair of terminals 24 and 25 is supplied to a full wave rectifier 30 which includes diodes D 1 , D 2 , D 3 and D 4 . The anode of diode D 1 and cathode of diode D 2 are connected to terminal 24. The anode of diode D 3 and cathode of diode D 4 are connected to terminal 25. The output of rectifier 30 (i.e. rectified a.c. signal) at a pair of output terminals 31 and 32 is supplied to a boost converter 40. The cathodes of diodes D 1 and D 3 are connected to terminal 31. The cathodes of diodes D 2 and D 4 are connected to terminal 32. Converter 40 boosts the magnitude of the rectified A.C. signal supplied by rectifier 30 and produces at a pair of output terminals 41 and 42 a regulated D.C. voltage supply. Boost converter 40 includes a choke L 3 , a diode D 5 the anode of which is connected to one end of choke L 3 . The other end of choke L 3 is connected to output terminal 31 of rectifier 30. The output of boost converter 40 at output terminals 41, 42 is applied across an electrolytic capacitor C E , one end of which is connected to the cathode of diode D 5 . A transistor (switch) Q 1 is connected to the junction between choke L 1 , and the anode of diode D 5 . The other end of transistor Q 1 is connected to the junction between the other end of capacitor C E , output terminal 32 of rectifier 30 and output terminal 42. A preconditioner control 50, which is powered by a D.C. supply voltage V, controls the switching duration and frequency of transistor Q 1 . Preconditioner control 50 is preferably, but not limited to, a Motorola MC33261 Power Factor Controller Integrated Circuit. Transistor Q 1 is preferably a MOSFET, the gate of which is connected to preconditioner control 50. Rectifier 30 and boost converter 40, including preconditioner control 50, form a preconditioner 80 for ballast circuit 20. Output terminals 41 and 42 of boost converter 40 serve as the output for preconditioner 80 across which a regulated D.C. voltage is produced. A lamp drive 90, which is supplied with the regulated D.C. voltage outputted by preconditioner 80, includes a half bridge inverter having a level shifter 60 and a half-bridge drive 70. The half bridge inverter includes a pair of transistors Q 6 and Q 7 , which serve as switches, a pair of capacitors C 5 and C 6 and a transformer T 1 . Half-bridge drive 70 produces a square wave driving signal to drive transistor Q 7 and has a 50-50 duty cycle. Level shifter 60 inverts the driving signal supplied to transistor Q 7 for driving transistor Q 6 . The driving signals produced by level shifter 60 and half-bridge drive 70 are approximately 180° out of phase with each other so as to prevent conduction of transistors Q 6 and Q 7 at the same time, respectively. A source S of transistor Q 6 and one end of level shifter 60 are connected to output terminal 41 of boost converter 40. A drain D of transistor Q 6 is connected to a terminal A. The other end of level shifter 60, one end of half-bridge drive 70 and a source S of transistor Q 7 are also are connected to terminal A. The other end of half-bridge drive 70 and a drain D of transistor Q 7 are connected to output terminal 42 of boost converter 40. Capacitor C 5 is connected at one end to output terminal 41. The other end of capacitor C 5 and one end of capacitor C 6 are connected to a terminal B. The other end of capacitor C 6 is connected to output terminal 42. A primary winding T p of transformer T 1 is connected to terminals A and B. A secondary winding T S is connected at one end to an inductor L 7 , the latter which generally represents either the leakage inductance of transformer T 1 or a discrete choke. Connected to the other end of inductor L 7 , is one end of a capacitor C 10 and one end of a lamp load LL. Lamp load LL can include any combination of lamps and is shown, but not limited to, the series combination of two fluorescent lamps LL 1 and LL 2 . The other ends of capacitor C 10 and lamp load LL are connected to the other end of secondary winding T s . The turns ratio between primary winding T p and secondary winding T s of transformer T 1 is N p /N s . Transformer T 1 electrically isolates lamp load LL from the output voltage produced by preconditioner 80 and provides sufficient open circuit voltage during pre-ignition to ignite lamp load LL. The inductance of inductor L 7 is based on the desired current flow through lamp load LL once the latter has ignited and is in its steady-state mode of operation. The DC voltage across each capacitor C 5 and capacitor C 6 is approximately half the output voltage of preconditioner 80. The waveforms shown in Figs. 4(a), 4(b), 4(c) and 4(d) produced by ballast circuit 20 are based on turns ratio N s /N p of about 1.5, inductor L 7 of approximately 4.3 millihenries, capacitor C 10 of about 1.2 nanofarads and capacitors C 3 and C 4 of about 0.33 microfarads, nominally rated at 630 volts. Both lamp LL 1 , and lamp LL 2 are 40 watt low pressure mercury vapor tubular fluorescent lamps. The fundamental frequency of the square wave produced by the half-bridge inverter is approximately 28kHz. The resonant frequency of inductor L 7 and capacitor C 10 is approximately 70kHz, that is, approximately 2.5 times fundamental frequency f 1 . During pre-ignition of lamp load LL, the output of the half-bridge inverter, which is across terminals A-B, forms a substantially square wave voltage train. Inductor L 7 and capacitor C 10 form an L-C series connected circuit. During pre-ignition, lamp load LL appears as a substantially open circuit (i.e. no load condition) drawing substantially no power expect for filament heating (assuming lamps LL 1 and LL 2 are fluorescent lamps of, for example, the rapid-start type). Fig. 4(a) illustrates a voltage V AB , that is, between terminals A and B. Voltage V AB is square wave voltage train which is applied across primary winding T p varying between approximately +240 volts and -240 volts during no load conditions. Fig. 4(b) illustrates current I PRI flowing through primary winding T p during no load conditions, that is, prior to ignition of lamp load LL and having a peak value of approximately ± 400 milliamperes. Once lamp load LL is ignited and is in its steady-state operation, current I PRI flowing through primary winding T p , as shown in Fig. 4(c), has a somewhat sinusoidal wave shape with a peak value of approximately ± 800 milliamperes. Capacitor C 10 serves to smooth this somewhat sinusoidal current waveform resulting in a substantially sinusoidal lamp current I LAMP as shown in FIG. 4(d) having a peak value of approximately ± 380 milliamperes. Inductor L 7 serves as the lamp current ballasting element. Capacitor C 10 , which is placed across lamp load LL, provides a more sinusoidal open circuit voltage and keeps total half bridge current inductive while also lowering higher harmonic content of current flowing through lamp load LL. Inductor L 7 and capacitor C 10 together form a series connected L-C output circuit. The value for capacitor C 10 is chosen such that safe open circuit operation is provided, that is, within the range of resonant frequencies defined by eq. 8. Accordingly, no additional circuits to protect lamp drive circuit 90 are required. When ballast circuit 20 is first turned on, prior to the voltage being boosted by preconditioner 80, the input voltage of approximately 277 volts results in a square wave voltage of approximately 390 volts peak to peak being applied across primary winding T p of transformer T 1 which is stepped up to approximately 570 volts peak to peak across secondary winding T s . During this time the lamp cathodes are heated. After approximately 0.5 seconds, preconditioner 80 turns ON resulting in a regulated D.C. voltage of approximately 480 volts across output terminals 41, 42 of boost converter 40 and a voltage of approximately 700 volts peak to peak across secondary winding T s , the latter of which is sufficient for igniting lamp load LL. Once lamp load LL is ignited (i.e. during steady-state lamp operation), the lamp voltage (i.e. voltage across lamp load LL) drops to approximately ± 300 volts peak with the remainder of the secondary winding T S output voltage across inductor L 7 . The number of and connections between the lamps within lamp load LL can be varied as desired with the value of inductor L 7 being chosen so as to provide the desired lamp current I LAMP during steady-state operation of lamp load LL. Referring again to Fig. 3, the rectified AC (i.e. pulsating DC) signal supplied to preconditioner 80 from diode bridge rectifier 30 is boosted in magnitude by choke L 3 , and diode D 5 to charge capacitors C E , C 5 and C 6 . In Fig. 3, capacitor C E is separate from capacitors C 5 and C 6 , capacitor C E being a large electrolytic capacitor in the range of 5 to 100 microfarads. Capacitors C 5 and C 6 are high frequency bridge capacitors. Since capacitor C E is in parallel with the series combination of capacitors C 5 and C 6 , these three capacitors can be reconfigured as capacitors C 5 ' and C 6 '. Preconditioner 80 is an up-converter and boosts the rectified AC input voltage as follows. When transistor Q 6 (which serves as a switch) is closed, choke L3 is short circuited to ground. Current flows through choke L 3 . Transistor Q 1 is then opened (turned OFF). Choke L 3 with transistor Q 1 open transfers stored energy through diode D 5 into capacitor C E . The amount of energy transferred to capacitor C E is based on the time during which transistor Q 1 is turned ON, that is, based on the frequency and duration of the driving signal supplied to the gate of transistor Q 1 by the preconditioner control 50. Asynchronous operation of transistor Q 1 with respect to voltage V LN results. Choke L 3 operates in a discontinuous mode, that is, the current through choke L 3 during each cycle is reduced to substantially zero before a new cycle is initiated. The frequency at which transistor Q 1 is turned ON and OFF is varied by preconditioner control 50 so that the peak current through choke L 3 is kept constant. Transistors Q 6 and Q 7 have internal diodes (not shown). These diodes, which can either be internal or external to the transistors, permit inductive currents to flow through transistors Q 6 and Q 7 at the initial turn ON and turn OFF of transistors Q 6 and Q 7 . Preferably, capacitors C 5 and C 6 are electrolytic capacitors having a pair of discharge resistors in parallel, respectively. Transformer T 1 is a leakage transformer, that is, having a leakage inductor of inductance LM which serves as the ballast for lamp load LL (i.e. to limit steady state current flow through the lamp load). Alternatively, when transformer T 1 has little or no leakage inductance an external inductor of inductance L M is required for ballast purposes. Transformer T 1 has a main secondary winding T M . A resonant capacitor C 10 is in series with inductor L 7 and reflects back to the primary winding of transformer T 1 as a series LC combination across the half-bridge inverter. As now can be readily appreciated, by maintaining the fundamental sinusoidal frequency f 1 well below resonant frequency f 0 of the series L-C output circuit, the undesirable and unsafe high voltages and current levels produced in conventional ballast circuits during pre-ignition of lamp load LL are avoided. More particularly, by choosing the values of inductor L 7 and capacitor C 10 such that their resonant frequency f 0 is defined as described hereinbefore, the voltage level across inductor L 7 and capacitor C 10 and current flow therethrough will be maintained at levels far below conventional ballast output circuits during pre-ignition of lamp load LL. By not requiring the combination of inductor L 7 and capacitor C 10 to be operated at its resonant frequency f 0 during pre-ignition of lamp load LL, the value of capacitor C 10 can be significantly reduced. For example, conventional values for capacitor C 10 range from about a nominal value of 6.8 nanofarads to about a nominal value of 9.2 nanofarads. In accordance with the invention, however, capacitor C 10 can be reduced in value by approximately one-fourth to one-sixth (e.g. to approximately 1.2 nanofarads). Consequently, a far smaller, less expensive capacitor C 10 is required reducing the manufacturing cost and space requirements of the ballast output circuit. The reduced value of capacitor C 10 results on top of this in substantially all current flowing through lamp load LL with relatively little current flowing through capacitor C 2 . Power requirements for the ballast circuit can be reduced and/or less costly wiring (higher resistance) can be used in the series connected L-C ballast output circuit while maintaining the same power requirements as in a conventional ballast output circuit. In other words, a less costly and/or more efficient ballast with smaller space requirements is provided by the present invention. Preferably, resonant frequency f 0 should range from approximately 2.3 to 2.6 times fundamental frequency f 1 of the square wave generated by the square wave generator. Consequently, stray inductances and the like which may be difficult to account for will not increase the overall inductance. Resonant frequency f 0 will not approach third harmonic frequency 3f 1 . Unsafe operation (i.e., resonant operation of the series L-C output circuit) of ballast circuit 20 is prevented. Generally, in calculating the inductance of inductor L 7 for determining resonant frequency f 0 , the leakage inductance of transformer T 1 or inductance of the discrete choke used for inductor L 7 is far greater than the stray inductance or other inductances within ballast circuit 20. Therefore, as a first order approximation, the inductance of inductor L 7 can be used without taking into account stray inductances and the like in determining the resonant frequency f 0 . For a tightly wound transformer T 1 in which very little or an insufficient amount of leakage inductance exists, a discrete inductor will be required to serve as the ballasting element for lamp load LL (i.e., to control the lamp current I LAMP ). As now can be readily appreciated, the generated voltage (i.e. voltage E of Fig. 1 and voltage V A-B of Fig. 4(a)) is at a frequency which is far less than the resonant frequency of the series connected L-C circuit and therefore provides safe open circuit (pre-ignition) voltages and current levels. The frequency of this generated signal need not be changed following pre-ignition since it is never at or near resonant frequency f 0 of the series connected L-C circuit. Feedback circuitry for sensing ignition of lamp load LL for switching to a different steady-state lamp operating frequency need not be provided. By eliminated the need to operate at resonant frequency f 0 of the series L-C circuit during pre-ignition of lamp load LL, the value and resulting size of the capacitor for the series connected L-C circuit can be far smaller than normally used in a conventional series connected L-C circuit.

Claims: A ballast circuit for generating a substantially rectangular driving signal sufficient to ignite a lamp load (LL), comprising: inductor means (L 7 ); capacitor means (C 10 ) serially connected to said inductor means (L 7 ); and generating means for applying the generated substantially rectangular driving signal to said serially connected inductor means (L 7 ) and capacitor means (C 10 ), said generated substantially rectangular driving signal having at least a fundamental frequency f 1 ; the inductor means (L 7 ) and capacitor means (C 10 ) having a resonant frequency f o , characterized in that for the fundamental frequency f 1 and the resonant frequency f o it holds: nf 1 < f o < (n+ 1)f 1 with n = an even integer. A ballast circuit of claim 1, characterized in that the generating means includes a half-bridge inverter. A ballast circuit of claim 1 or 2, wherein the resonant frequency f o is less than a third harmonic of said fundamental frequency f 1 . A ballast circuit of claim 1, 2 or 3, wherein the lamp load after ignition enters into a steady-state mode of operation in which current therethrough is maintained at a substantially constant level, characterized in that in the steady-state mode the generating means apply said generated signal to said serially connected inductor means and capacitor means. A ballast circuit of claim 1, 2, 3 or 4 wherein said lamp load is connected across the capacitor means. A ballast circuit of claim 1, 2, 3, 4 or 5, wherein the lamp load includes at least one fluorescent lamp.

Inventors: MATTAS CHARLES, MATTAS, CHARLES
Applicants: PHILIPS ELECTRONICS NV, PHILIPS ELECTRONICS N.V.
Application year: 1993
Main classification: H05B41
IPC codes: H05B41