Voltage converting circuit and multiphase clock generating circuit for driving the same

Abstract: A voltage converting circuit of the charge pump step-up type includes a first circuit means for charging each of first and second capacitors with the voltage of a voltage source at a first timing. A second circuit operates to serially connect the charged first capacitor between a positive electrode of the voltage source and a positive voltage output terminal at a second timing so that a positive voltage which is a double of the voltage source voltage, is supplied from positive voltage output terminal. A third circuit operates to the charged first and second capacitors in series between a ground terminal and a negative voltage output terminal at a third timing so that a negative voltage which is a double of the voltage source voltage, is supplied from the negative voltage output terminal. Since the positive voltage and the negative voltage are generated independently of each other, a voltage variation on one of the positive and negative voltage output terminals caused by an external load causes no voltage variation on the other of the positive and negative voltage output terminals.

Description: Background of the Invention Field of the invention The present invention relates to a voltage converting circuit and a multiphase clock generating circuit used for the same, and more specifically to a charge-pump type voltage converting circuit for generating a large positive and negative voltage from a single voltage source, and a multiphase clock generating circuit for driving the charge-pump type voltage converting circuit. Description of related art In the prior art, various voltage converting circuits and multiphase clock generating circuits used for the voltage converting circuits have been known. The voltage converting circuits can be classified into a step-up circuit and a step-down circuit. (1) Step-up Voltage Converting Circuit Conventionally, in order to obtain from a single voltage source a positive or negative output voltage having the magnitude larger than a voltage source voltage, the step-up voltage converting circuit has been widely used in a voltage supply circuit for RS-232C driver/receiver IC (integrated circuit) and others. Some typical examples of the conventional step-up voltage converting circuit are disclosed in US Patents 4,777,577, 4,897,774, 4,999,761, 4,807,104 and 4,812,961. The step-up voltage converting circuits shown in these US patents are configured to be operated with a 2-phase clock, and is constructed of a so-called switched capacitor circuit type. For example, the conventional step-up voltage converting circuit disclosed in Figure 1A of US Patent 4,777,577 includes a step-up circuit part for generating a voltage which is a double of a voltage source voltage so that a doubled voltage is outputted from a positive voltage output terminal 40, and an inverting circuit part for generating an inverted voltage having the same magnitude as the doubled voltage so that the inverted doubled voltage is outputted from a negative voltage output terminal 38. However, when the positive voltage output terminal 40 is connected to a large load which causes a voltage of positive voltage output terminal to drop, a voltage of a positive reservoir capacitor 22 and a voltage of an inverting capacitor 24 correspondingly drop. Namely, the absolute value of the voltage of the negative voltage output terminal 38 becomes small, with the result that another circuit connected to the negative voltage output terminal 38 becomes unable to maintain a stable operation. In addition, since variation in the voltage of the positive voltage output terminal 40 causes variation in the voltage of the negative voltage output terminal 38, if a voltage supply circuit is constituted of this step-up circuit, the variation of the output voltage becomes double, with the result that an overall system including the voltage supply circuit therein has a remarkably lowered reliability. Furthermore, the conventional step-up circuit can generate a pair of positive and negative voltages such as ±2V DD or ±3V DD , which have the same absolute value and which have the magnitude that is integer times the voltage source voltage. However, it is sometimes required to supply positive and negative biasing voltages having different absolute values, for example, +3V DD and -2V DD , as in a bias voltage generating circuit for a CCD (charge coupled device) driver IC. For this application, the conventional step-up circuit cannot be used. (2) Step-down Voltage Converting Circuit Conventionally, the step-down voltage converting circuit has been used for obtaining from a single voltage source a positive or negative output voltage having the magnitude smaller than a voltage source voltage. If this step-down circuit is incorporated on a printed circuit board, it is ordinary cases to use a three-terminal voltage regulator or a switching regulator using a solenoid. The three-terminal voltage regulator has to be implemented in a bipolar process, and has a large loss in its output stage transistor. On the other hand, the switching regulator has a loss smaller than that of the three-terminal regulator, but inevitably has a large scale since the solenoid must be incorporated. Under this circumstance, in the case of incorporating a step-down circuit in a CMOS (complementary metal-oxide-semiconductor) integrated circuit, a switched capacitor type step-down circuit has been used which is highly compatible with a CMOS integrated circuit manufacturing process and which has less loss. One typical example of the conventional switched capacitor type step-down circuit is shown in Journal of Japan Society of Electronics and Communication Engineers, 83/8, Vol. J66-C, No.8, pp576-583. This step-down circuit includes a reservoir capacitor connected between a positive voltage output terminal and a ground terminal, and a transfer capacitor having the same capacitance as that of the reservoir capacitor. During a first phase, these capacitors are connected in series between a positive electrode and a negative electrode of a voltage source, so that each of the capacitors is charged to a half of the voltage of the voltage source, and therefore, the half of the voltage source voltage is outputted from the positive voltage output terminal. During a second phase complementary to the first phase, the capacitors are separated from the voltage source but connected in parallel to each other, so that the voltage of the positive voltage output terminal is maintained at the half of the voltage source voltage. However, this conventional step-down circuit cannot give positive and negative voltages having their absolute value which is a half of the voltage source voltage. For example, it may be considered to add an inverting circuit which inverts the voltage on the positive voltage output terminal so as to supply a negative voltage having the same magnitude as the voltage of the positive voltage output terminal. This modification can surely give positive and negative output voltages, however, when the voltage of the positive voltage output terminal varies due to influence of an external load, the output voltage of the inverting circuit connected to the positive voltage output terminal correspondingly varies, with the result that an external circuit connected to the inverting circuit may cause a malfunction. In addition, if a voltage supply circuit is composed of the step-down circuit having the added inverting circuit, the voltage variation on the positive voltage output terminal directly becomes the voltage variation on a negative voltage output terminal, and therefore, the voltage variation of the voltage supply circuit is doubled. Accordingly, reliability of the voltage supply circuit remarkably lowers. (3) Multiphase Clock Generating Circuit Conventionally, multiphase clock generating circuits have been used with a switching-element containing circuit such as the switched capacitor step-up circuit, the switched capacitor step-down circuit and others, for the purpose of supplying timing clocks to switching elements for switch-over between an ON condition and an OFF condition of each switching element. In these cases, in order to ensure that switches to be on-off controlled at different phases are never simultaneously put in the ON condition, it is necessary to use timing clock signals which are different in phase and which never overlap each other. For example, the conventional step-up voltage converting circuit disclosed in Figure 1A of US Patent 4,777,577 needs a two-phase clock generator. In many cases, in addition, it becomes necessary to control the switching elements with clocks of three or more different phases, and therefore, a multiphase clock generating circuit for generating clocks of three or more different phases must be used. The conventional multiphase clock generating circuits having the above mentioned functions have required at least one 1/2 frequency divider composed of for example a D-type flipflop, in order to generate from a single input clock a pair of clock signals that do not overlap each other in phase. Accordingly, in order to generate an N-phase clock, at least N 1/2 frequency dividers (each composed of a D-type flipflop) are required. This is disadvantageous since the number of necessary elements inevitably becomes large and therefore since it needs a large chip area when it is implemented in an integrated circuit. Furthermore, since the multiphase clock is generated by action of the frequency division realized by the flipflops, a clock to be supplied to a multiphase clock generating circuit has to have a sufficiently high frequency. In this connection, a possible 3-phase clock generating circuit mainly composed of flipflops would require an input clock having the frequency which is four times the frequency of each of three clocks of different phases generated by the 3-phase clock generating circuit itself. As a result, a large amount of a pass-through current flows from the voltage source to the ground, and therefore, a consumed electric power is large. US-A-4,816,700 discloses a two-phase non-overlapping clock generator. Therein, an external timing source provides an input waveform of signal voltage transitions which occur at a first frequency, which is twice the frequency desired by the internal clock. A divider circuit divides the input waveform into a pair of intermediate waveforms. The intermediate waveforms and signal-voltage transitions of opposing polarity which occur at a second frequency that is one-half of the first frequency. The input waveforms and the intermediate waveforms are supplied to a driver circuit. The driver circuit utilizes the intermediate waveforms occurring at the divided-down frequency and the input waveform occurring at the system clock frequency to produce a pair of final waveforms. The final waveforms have signal voltage transitions of opposing polarity occurring at the divided-down frequency, but triggered by and in synchronism with the input waveform occurring at the external clock frequency. Summary of the Invention It is an object of the present invention to provide a multiphase clock generating circuit which can be constructed with using no flipflop in an initial stage, and which can therefore be realized with a reduced chip area. Present invention is defined in independent claim 1. Preferred embodiment is described in the dependent claim 2. According to an embodiment of the present invention, there is provided a multiphase clock generating circuit including a 2-phase clock generating circuit composed of a CMOS level shift circuit having a predetermined delay time and receiving a input clock and generating a non-inverted output clock and an inverted output clock of different phases, and pulse selection means receiving one of the non-inverted output clock and the inverted output clock and generating a plurality of output signals which have a phase different from each other and which has a frequency obtained by dividing the frequency of the received clock by the number of the output signals. The above and other objects, features and advantages of the present invention will be apparent from the following description of preferred embodiments of the invention with reference to the accompanying drawings. Brief Description of the Drawings Figure 1 is a circuit diagram of a first embodiment of the 3-phase clock generating circuit in accordance with the present invention; Figure 2 is a timing chart illustrating an operation of the 3-phase clock generating circuit shown in Figure 1. Figure 3 is a circuit diagram of a step up circuit associated with an embodiment of the 3-phase clock generating circuit in accordance with the present invention; Figure 4 is a circuit diagram of the 3-phase clock generating circuit used in the embodiment shown in Figure 3; and Figure 5 is a timing chart illustrating an operation of the 3-phase clock generating circuit shown in Figure 4. Description of the Preferred embodiments Referring to Figure 1, there is shown a circuit diagram of a first embodiment of the 3-phase clock generating circuit in accordance with the present invention. The 3-phase clock generating circuit shown in Figure 1 is mainly composed of a flipflop type COMS level shift circuit 101, a buffer 103 and a pulse selection circuit 109. The COMS level shift circuit 101 includes a clock input terminal 100 connected to an input of an inverter 112 and a gate of an N-channel transistor 114 having its source connected to a ground terminal 111. An output of the inverter 112 is connected to a gate of an N-channel transistor 113 having its source connected to the ground terminal 111. A drain of the N-channel transistor 113 is connected to an output terminal 117 of the COMS level shift circuit 101, a drain of a P-channel transistor 115 and a gate of a P-channel transistor 116. These P-channel transistors 115 and 116 have their source connected in common to a voltage supply terminal 110, V DD . A drain of the N-channel transistor 114 is connected to another output terminal 118 of the COMS level shift circuit 101, a gate of the P-channel transistor 115 and a drain of the P-channel transistor 116. The output terminal 117 is connected through the buffer 103 to an output terminal of 106 of the 3-phase clock generating circuit. The pulse selection circuit 109 includes a D-type flipflop 119 having a clock input C connected to the output terminal 118 of the COMS level shift circuit 101. A Q output of the D-type flipflop 119 is connected to a data input D of the D-type flipflop 119 itself and one input of an AND circuit 105. A Q output of the D-type flipflop 119 is connected to one input of an AND circuit 104. The other input of each of the AND circuits 104 and 105 is connected in common to the clock input C of the D-type flipflop 119. An output of the AND circuits 104 and 105 are connected to output terminals 107 and 108 of the 3-phase clock generating circuit, respectively. Here, since the Q output of the D-type flipflop 119 is connected to the data input D of the D-type flipflop 119 itself, the D-type flipflop 119 constitutes a 1/2 frequency divider circuit 102. Now, operation of the 3-phase clock generating circuit shown in Figure 1 will be described with reference to a timing chart of Figure 2 illustrating an operation of the 3-phase clock generating circuit shown in Figure 1, Figure 2 illustrate a change in time of each of the input clock CLK supplied to the clock input terminal 100, a pair of output signals C1 and C2 appearing on the output terminals 117 and 118 of the CMOS level shift circuit 101, the Q output and the Q output of the D-type flipflop 119, and output signals  1 ,  2 and  3 appearing on the output terminals 106, 107 and 108. As shown in Figure 2, when the input clock CLK are at a low level, the N-channel transistors 113 and 114 are on and off, respectively, and therefore, the output signals C1 and C2 are at a low level and at a high level, respectively. When the input clock CLK changes to the high level, the N-channel transistors 113 and 114 are turned off and on, respectively, and therefore, the output signal C2 becomes the low level. Since the output signal C2 is applied to the gate of the P-channel transistor 115, after the output signal C2 becomes the low level, the P-channel transistor 115 is turned on after a constant delay time which is determined by an ON resistance of the N-channel transistor 114 and a gate capacitance of the P-channel transistor 115. The output signal C1 becomes the high level upon the turning-on of the P-channel transistor 115. Furthermore, as shown in Figure 2, when the input clock CLK changes to the low level, the N-channel transistors 113 and 114 are turned on and off, respectively, and therefore, the output signal C1 becomes the low level. Since the output signal C1 is applied to the gate of the P-channel transistor 116, after the output signal C1 becomes the low level, the P-channel transistor 116 is turned on after a constant delay time which is determined by an ON resistance of the N-channel transistor 113 and a gate capacitance of the P-channel transistor 116. The output signal C2 becomes the high level upon the turning-on of the P-channel transistor 116. As will be apparent from the above and from Figure 2, the output signals C1 and C2 constitute two different phase clock signals having a high level period that does not overlap each other. In this case, the N-channel transistors 113 and 114 are designed to have a current driving capacity larger than that of the P-channel transistors 115 and 116, in order that, in operation, the output signals C1 and C2 of the output terminals 117 and 118 can be surely inverted. For example, when the input clock signal CLK has changed from the low level to the high level, the N-channel transistor 114 is immediately turned on. During the delay time in which the P-channel transistor 116 remains on, the N-channel transistor 114 must flow a current larger than the drain current of the P-channel transistor 116 in order to invert the output signal C2 to the low level. Here, assuming that the input clock signal CLK has an amplitude of 5 V and the voltage V DD of the voltage supply terminal 110 is 10 V, the gate voltage of the N-channel transistor 114 is 5 V at this time, and the gate voltage of the P-channel transistor 116, which is a double of the gate voltage of the N-channel transistor 114. Accordingly, since the current drive capacity of a MOS (metal-oxide-semiconductor) field effect transistor is in proportion to a square of the gate voltage, the gate width/gate length ratio of the N-channel transistor 114 is set to be larger than a value which is four times the gate width/gate length ratio of the P-channel transistor 116. Actually, since mobility of electrons is generally larger than mobility of holes, it is sufficient if the gate width/gate length ratio of the N-channel transistor 114 is four times as larger as the gate width/gate length ratio of the P-channel transistor 116. The output signal C1 is supplied through the buffer 103 from the output terminal 106, as the output signal  1 of the 3-phase clock generating circuit, and the output signal C2 is supplied to the clock input C of the 1/2 frequency divider circuit 102. Thus, the Q output of the 1/2 frequency divider circuit 102 generates the output signal Q which is flipped each time the output signal C2 is brought to the high level. The Q output of the 1/2 frequency divider circuit 102 generates the output signal Q complementary to the output signal Q as shown in Figure 2. These output signals Q and Q are supplied to the AND circuits 104 and 105, respectively, so that the logical product of the output signals Q and Q and the output signal C2 are supplied from the output terminals 107 and 108 as the output signals  2 and  3 , respectively. Accordingly, the clock signals  1 ,  2 and  3 of different phases having a high level period that never overlap each other, can be obtained from the output terminals 106, 107 and 108. These clock signals are sequentially brought to the high level in the order of  1 ,  2 ,  1 , and  3 in time. The four clock signals  1 ,  2 ,  1 , and  3 constitute one cycle, and therefore, are repeated cyclicly. The 3-phase clock generating circuit shown in Figure 1 uses only one 1/2 frequency divider circuit formed of the D-type flipflop, which needs a large chip area. Accordingly, the whole of the 3-phase clock generating circuit shown in Figure 1 requires only a reduced chip area, and therefore, the 3-phase clock generating circuit shown in Figure 1 can be easily realized on an integrated circuit. In addition, the clocks of two different phases that never overlap each other, are generated by utilizing the gate delay time in the CMOS level shift circuit. Therefore, it is not required that the frequency of the input clocks applied to the 3-phase clock generating circuit is sufficiently higher than that of the output clocks of the 3-phase clock generating circuit. Namely, as seen from Figure 2, it is sufficient if the frequency of the input clock signal CLK is a double of the frequency of the output clock signals  2 and  3 of the 3-phase clock generating circuit. The frequency of the input clock signal CLK is substantially the same as the frequency of the output clock signal  1 of the 3-phase clock generating circuit. Accordingly, an oscillator circuit (not shown) for supplying the input clock CLK can be easily constructed. Furthermore, since the input clock CLK does not require a high frequency, the pass-through current flowing through the MOS transistors from the voltage supply terminal to the ground terminal within the 3-phase clock generating circuit, can be remarkably reduced, with the result that the consumed electric power becomes correspondingly small. The above mentioned embodiment of the multiphase clock generating circuit is of the 3-phase clock generating circuit. However, the multiphase clock generating circuit of four phases of more can be similarly constructed. For example, an additional pulse selection circuit is connected to one of the output terminals of the 3-phase clock generating circuit, so that the additional pulse selection circuit generates two clock signals which never overlap each other and which have a frequency corresponding to a half of the frequency of the output clock outputted form the output terminal of the 3-phase clock generating circuit connected to the additional pulse selection circuit. Thus, a circuit composed of the 3-phase clock generating circuit and the additional pulse selection circuit functions as a four-phase clock generating circuit as a whole. Accordingly, the multiphase clock generating circuit can be constructed by connecting a pulse generating circuit, and the arrangement of the multiphase clock composed of the whole of output pulses can be determined by selecting output terminals of newly added pulse selection circuits. For example, if an additional pulse selection circuit is connected to the output terminal 108 of the 3-phase clock generating circuit shown in Figure 8 so that a pair of output signals  4 and  5 are generated by the additional pulse selection circuit, there can be obtained a four-phase clock generating circuit which generates four different phase clocks  1 ,  2 ,  4 and  5 which never overlap each other and which are cyclicly brought to the high level in the order of  1 ,  2 ,  1 ,  4 ,  1 ,  2 ,  1 , and  5 , which constitute one cycle. Referring to Figure 3, there is shown a circuit diagram of an embodiment of a step-up circuit invention associated with the 3-phase clock generating circuit in accordance with the present invention. The step-up circuit shown in Figure 10 is a step-up circuit in which the switches are realized in a CMOS transistor technology. The positive step-up switches in the step-up circuit are constituted of positive step-up P-channel transistors 133 and 134, respectively. The negative step-up switches in the step-up circuit are constituted of negative step-up N-channel transistors 135, 136 and 137, respectively. The charging switches are constituted of charging P-channel transistors 129 and 130, respectively. The charging switches are constituted of charging N-channel transistors 131 and 132, respectively. As show in Figure 3, respective gates of these N-channel and P-channel transistors are supplied with the timing clock signals  1 and  3 outputted from a 3-phase clock generating circuit 139 and inverted signals of timing clock signals  1 and  2 obtained through inverters 140 and 141. In addition, the voltage source and the capacitors correspond to a voltage source 120 and capacitors 122, 125, 127 and 128 of Figure 3, respectively. The output terminals correspond to output terminals 123 and 126, respectively. In the above mentioned step-up circuit shown in Figure 3, the switches connected to a high level voltage supply terminal 121 or the positive voltage output terminal 123 are formed of the P-channel transistor, and the switches connected to a ground terminal 124 or the negative voltage output terminal 126 are formed of the N-channel transistor. Therefore, since an on resistance of these transistors can be made small by applying a sufficiently large gate voltage, the step-up efficiency can be elevated, and the loss can be decreased with a reduced consumed power. In addition, the 3-phase clock generating circuit 139 are driven with +2V DD and -2V DD as a voltage supply voltage, so that the output tining signals  1 ,  2 and  3 for switching the N-channel and P-channel transistors have a sufficient amplitude between a positive voltage and a negative voltage. Referring to Figure 4, there is shown a circuit diagram of the 3-phase clock generating circuit 139 used in the embodiment shown in Figure 3. In this embodiment, it is assumed that V DD is 5 V. A clock signal CLK having an amplitude from 0 V to 5 V is applied to a clock input terminal 150 corresponding a clock input 138 of Figure 3. This clock signal CLK is inverted by an inverter 164 composed of a P-channel transistor 165 and an N-channel transistor 166. The clock signal CLK and its inverted signal are supplied to a first level shift circuit 167 which is composed of P-channel transistors 168 and 169 and N-channel transistors 170 and 171 connected as shown in Figure 4. Namely, the clock signal CLK and its inverted signal are converted into a pair of complementary signals having an amplitude from -10 V to 5 V Furthermore, the pair of complementary signals are supplied to a second level shift circuit 172 which is composed of N-channel transistors 173 and 174 and P-channel transistors 175 and 176 connected as shown in Figure 4. Namely, the pair of complementary signals are converted into a pair of clock signals having an amplitude from -10 V to +10 V, which are supplied to a pair of inverters 153 and 159, respectively. An output signal C1 of the inverter 153 is supplied from an output terminal 156 as a first timing signal  1 . An output signal C2 of the inverter 159 is supplied to a pulse selection circuit 149 constructed similarly to the pulse selection circuit 109 shown in Figure 1. Thus, second and third signals  2 and  3 are supplied from output terminals 157 and 158, respectively. Figure 5 is a timing chart illustrating the change in time of the input clock CLK and the first, second and third timing signals  1 ,  2 and  3 in the 3-phase clock generating circuit shown in Figure 4. Since the CMOS level shift circuit 151 is constituted by using two CMOS level shift circuits which are similar to the CMOS level shift circuit used in the 3-phase clock generating circuit of Figure 1 and which are coupled to form a two-stage cascaded structure, the timing signals having an amplitude from -2V DD to +2V DD can be obtained. Therefore, it is possible to drive the step-up circuit of the CMOS transistor structure shown in Figure 3. As will be apparent from the above, the voltage converting circuit in accordance with the present application is configured so that a positive voltage and a negative voltage are generated independently of each other, by using the switches on-off controlled by three different phase clock signals. Accordingly, a voltage variation on one of the positive and negative voltage output terminals caused by an external load causes no voltage variation on the other of the positive and negative voltage output terminals. Therefore, an overall system including the voltage converting circuit in accordance with the present application as a voltage supply circuit, can have an elevated reliability. In addition, the step-up circuit is capable of generating positive and negative voltages having different absolute values. The step-down circuit is capable of generating not only a positive voltage but also a negative voltage. The multiphase clock generating circuit in accordance with the present application is configured so that an initial stage for generating from a single input clock a pair of different phase clocks that do not overlap each other, is constituted of a CMOS level shift circuit, not by the D-type flipflop which needs a large chip area. Accordingly, the whole of the 3-phase clock generating circuit can be realized with a reduced chip area. In addition, since the input clock signal is not frequency-divided by by a 1/2 frequency-divider, the frequency of the input clock signal CLK can be set at a low value. This is very effective in reducing the pass-through current and hence the consumed electric power.

Claims: A multiphase clock generating circuit providing a plurality of non-overlapping clock signals ( 1 ,  2 ,  3 ), comprising: a CMOS level shift circuit (101, 151) having a predetermined delay time and receiving an input clock (CLK) and generating a non-inverted output clock (C1) and an inverted output clock (C2) of different phases, and pulse selection means (109, 149) receiving one of said non-inverted output clock (C1) and said inverted output clock (C2) and generating a plurality of output signals which have a phase different from each other and which have a frequency obtained by dividing the frequency of the received clock by the number of said output signals, wherein the other of said non-inverted output clock (C1) and said inverted output clock (C2) being outputted as one of said non-overlapping clock signals ( 1 ,  2 ,  3 ). A multiphase clock generating circuit as claimed in claim 1 wherein the other of said non-inverted output clock (C1) and said inverted output clock (C2) is outputted as a first non-overlapping clock signal ( 1 ), and said pulse selection means (109, 149) including a 1/2 frequency dividing circuit (102, 152) receiving either said one of said non-inverted output clock (C1) or and said inverted output clock (C2) for generating both a non-inverted signal (Q) and an inverted signal ( Q ), a first AND gate (104, 154) receiving said non-inverted signal (Q) and either said non-inverted output clock (C1) or said inverted output clock (C2) for generating a second non-overlapping clock signal ( 2 ), and a second AND gate (105, 155) receiving said inverted signal ( Q ) and either said non-inverted output clock (C1) or said inverted output clock (C2) for generating a third non-overlapping clock signal ( 3 ).

Inventors: TAMAGAWA AKIO, TAMAGAWA, AKIO
Applicants: NEC ELECTRONICS CORP, NEC ELECTRONICS CORPORATION
Application year: 1993
Main classification: G06F1
IPC codes: H03K3, H02M3