Low supply voltage, band-gap voltage reference

Abstract: A voltage, replica of the difference between two dissimilar base-emitter voltages in the form of an intrinsic input offset voltage of a differential input pair of transistors of a noninverting, buffer-configured operational amplifier, is summed with a pre-established fraction of a base-emitter voltage, to produce a voltage reference without thermal drift of a level that can be as low as few 10mV. The intrinsic input offset voltage is controlled by a local feedback loop acting on the bias current that is forced through the input pair of transistors that may be realized with a certain area ratio. The relatively simple circuit is useful in battery operated, low supply voltage, systems.

Description: FIELD OF THE INVENTION The present invention relates to a method and a circuit for generating a reference voltage without thermal drift and of relatively low value, markedly lower than the voltage of a base-emitter junction (Vbe). BACKGROUND OF THE INVENTION In many systems and particularly in monolithically integrated systems, it is necessary to implement voltage references, that is circuits capable of generating a stable reference voltage, free of thermal drift. Commonly this is achieved by employing a so-called band-gap circuit. A band-gap circuit produces a voltage corresponding to the sum of one or several base-emitter voltages (Vbe), as of common bipolar junction transistors, and of a voltage proportional to the difference between two different base-emitter voltages, suitably amplified by a certain amplification factor K, so as to make the amplified difference voltage comparable with the voltage of one or several base-emitter junctions, in order to produce a desired reference voltage given by: Vref = Vbe + KΔVbe, where K > 1 The ΔVbe term that is employed for compensating the thermal drift of a certain sign of the particular Vbe or sum of Vbe used, may suitably assume a thermal coefficient of opposite sign of the thermal coefficient of the Vbe term used. Therefore, the resulting reference voltage Vref that is produced may be stable in terms of temperature variations. US-A-4,524,318 discloses a circuit for generating temperature stable reference voltage by summing a base-emitter voltage and a difference between two base-emitter voltages, multiplied by a constant greater than one in order to make comparable the two terms of the sum, wherein the difference between two base-emitter voltages is derived from a pair of transistors of different emitter-area constituting a differential stage. Commonly band-gap circuits produce a temperature compensated voltage Vref greater or equal to about 1.2V. On the other hand, in systems designed for operating with relatively low supply voltages, for example in battery powered portable instruments and apparatuses, the supply voltage may be relatively low, for example in the vicinity of 1.0V. This makes a correct operation of a normal band-gap circuit impossible. Recently, a band-gap reference voltage generating circuit has been proposed which is capable of providing a regulated voltage of relatively low level, in the vicinity of 200mV, which may be adjusted upward to higher levels. This makes the voltage reference circuit suitable also in battery powered systems with a supply voltage of just 1V. The circuit is described in the article entitled: "A Curvature-Corrected Low-Voltage Bandgap Reference", by Gerard C. M: Meijer, et al., IEEE Journal of Solid State Circuits, Vol. 28, No. 6, June 93, the content of which is herein incorporated by express reference. Such a known circuit adopts a compensating system of the nonlinearity of the temperature characteristics of a base-emitter junction (Vbe). Basically, the circuit employs a first circuit block for generating a current proportional to the absolute temperature (PTAT) and a second circuit block capable of generating a current proportional to a Vbe, plus a correction current for compensating the nonlinearity of the temperature coefficient of the Vbe. Thereafter, the sum of the two currents is converted to a voltage signal which is amplified by an output buffer. The circuit is relatively complex and generates a stabilized reference voltage of about 200mV, with a supply voltage that may be as low as about 1V. There remains a need or utility of a circuit capable of generating a reference voltage of a relatively low value, in the order of few 10mV free of thermal drift and that be relatively simple to realize. This objective is fully met by the method and the circuit object of the present invention. Basically, the method of the invention rests on the generation of a stabilized voltage in the form of a sum of a voltage equivalent to the difference between two different base-emitter voltages, which is advantageously represented by a suitably controlled intrinsic offset voltage of a pair of transistors that constitute an input differential stage of a buffer-configured, operational amplifier, and a pre-established fraction of a base-emitter junction voltage. A subdivision of a Vbe voltage is implemented by mirroring, in a certain ratio, a current proportional to a Vbe voltage and by converting the fractionary mirrored current into a fractionary Vbe voltage on a resistance. The voltage difference between two different base-emitter junction voltages to be summed with the fractionary portion of a Vbe voltage, in order to compensate in terms of temperature the resulting voltage sum, is obtained in the form of an intrinsic offset voltage, controlled through a local feedback loop, of a differential pair of transistors that form an input stage of an operational amplifier that practically works as an output buffer of the stabilized voltage produced by the circuit. The stabilized voltage sum that can be generated by the circuit may be of several 10mV and may be freely scaled-down by the use of a resistive voltage divider. The circuit may be powered with a voltage of about 1V, without jeopardizing its operation. Therefore the circuit is particularly useful in low voltage, battery powered systems. The different aspects and advantages of the invention will be more easily understood through the following description of important embodiments and by referring to the attached drawings, wherein: Figure 1 is a diagram of a circuit for generating a reference voltage, according to the present invention; Figure 2 is a partial, simplified circuit diagram of the circuit of Fig. 1, which enphasizes some essential aspects of the circuit of the invention; Figure 3 shows a voltage-temperature characteristic of a circuit made according to the present invention. With reference to Fig. 1, the portion on the circuit of the left-hand side of the node V1 will be commented first. The bipolar junction transistor (BJT) Q2 generates a current I given by the ratio between its base-emitter voltage Vbe and the resistance R1: I=Vbe Q2 /R1. A suitable start-up circuit may comprise, as shown, a current generator I1, which in practice may be constituted by a transistor Q0 of an appropriate size. As a matter of fact, a so-called start-up circuit is necessary in order to ensure that, at the turn-on instant, the local loop reach a self-sustaining condition. By first approximation, the base current of the BJT Q2 under equilibrium conditions, will be given by: Ic Q2 =I1. This condition will then be maintained stable by the local feedback loop. However, at the turn-on instant, Q2 is still off and will turn-on only when Ic Q4 =Vbe Q2 /R1. Therefore, the collector voltage of Q2 will tend to drop until Q1 (which triggers the feedback) turns on thus supplying a current to Q3, which current, mirrored by Q4, drives the base of Q2. This driving current of the BJT Q2 will continue to increase until the following relationships verify: Ic Q1 =Ic Q3 =Ic Q4 =Vbe Q2 /R2 The transistor Q5, having the same area of Q1, will also conduct a current given by: Ic Q5 =Vbe Q2 /R2, which will be forced on R2, thus producing the voltage signal V1. The resistances RE Q1 and RE Q5 serve for degenerating the respective current generators Q1 and Q5. Therefore, a fraction of the Vbe Q2 voltage, given by V1=Vbe Q2 *R2/R1 is obtained, wherein the coefficient K=R2/R1 may be fixed according to needs. Such a fractionary portion V1 of a base-emitter junction voltage (Vbe Q2 ), as shown in Figures 1 and 2, is applied to the base of a first transistor Q6 of a differential input pair composed of Q6 and Q7, which practically represents a noninverting input of an operational amplifier, configured as a noninverting buffer. The inverting input of the amplifier, represented by the base node of the Q7 transistor of the differential input pair, is connected to an intermediate node (V2) of a resistive voltage divider R7-R6 of the output voltage produced by the operational amplifier. An analysis of the operation of the circuit of the invention will be rendered more easily, by momentarily referring to the partial and simplified circuit diagram of Fig. 2. Concisely, the transistor pair, Q6-Q7, and the generator I2 form a differential input stage. The transistor Q10 and its load, constituted by a diode-configured transistor Q11 and by a resistance R5, constitute an amplifying stage, while the transistor Q12 constitutes an output stage of the operational amplifier. The amplifier is configured as a noninverting buffer by means of a feedback line constituted by the resistance R7, connected between the output node (Vout) of the amplifier and its inverting input, that is the base node of the transistor Q7 of the input differential pair, and by the resistance R6 connected between the noninverting input and ground. As already said above, the effectiveness of the voltage reference circuit resides on the fact that the thermal drift of a certain sign of the fractionary portion V1 of a Vbe voltage, be counterbalanced by a thermal drift of opposite sign of a voltage difference between two different Vbe voltages, in order to ensure that the resulting sum voltage (V2) has a substantially null temperature coefficient (or thermal drift). As a voltage difference between two different Vbes, to be summed with the fractionary voltage V1 in order to obtain a resulting sum voltage that is temperature stable, is advantageously used an intrinsic offset voltage of the input pair of transistors Q6 and Q7 that form the input differential stage of the operational amplifier. A certain intrinsic offset voltage may be created by appropriately making the two transistors Q6 and Q7 that form the input differential pair with different emitter areas. Moreover, the offset voltage is controlled through a dedicated control loop of the bias current that is forced through the input pair of transistors. By referring to the functional diagram of Fig. 2, such a control loop (local feedback) of the bias current forced through the input pair of transistors Q6 and Q7 is implemented by the transistors Q8 and Q9, by the respective current generators I3 and I4 and by the resistances R3A and R3B. By assuming negligeable (in first approximation) the base current absorbed by the transistor Q10 and, for example, realizing Q8 and Q9 with identical emitter areas and forcing through Q8 and Q9 an identical current by the use of identical generators I3 and I4, each capable of generating a current I, the transistors Q8 and Q9 will assume an identical Vbe. This, coupled with the fact that the respective bases are connected in common, implies that the emitter voltage of Q8 be identical to the emitter voltage of Q9. This in turn permits to establish a certain current I b through R3B and a certain current I a through R3A, which will have the same ratio (for example 1:2) of the value of the resistances R3B and R3A. As may be observed, the current I a that flows through R3A contains also a contribution coming from the collector of Q6. Moreover, by assuming that the current generator I2 of the input differential stage generates a current nI, it is evident that the control loop fixes a certain collector current of the input transistor Q6 and, as a consequence, the collector current of the other transistor Q7 of the input differential pair will also be fixed by the local feedback loop, at the value given by the following expression: IC Q7 = nI - I = (n-1)I By applying Kirchoff law, V2 = V1 + Vbe Q6 - Vbe Q7 = Vbe R2 R1 + KT q ln I c6 Ae Q6 I S KT q ln I c7 Ae Q7 I S = Vbe R2 R1 + KT q ln I (n-1)I Ae Q7 Ae Q6 It may observed from the above indicated expresions, that the difference between the respective Vbe voltages of the transistors Q6 and Q7 may, in function of the ratio between their respective emitter areas, Ae Q7 /Ae Q6 , assume a temperature coefficient that can be either negative or positive and suitable for compensating the temperature coefficient of a certain sign possessed by the fractionary voltage V1. In the depicted example, the fractionary voltage V1 has a negative temperature coefficient and therefore the intrinsic offset voltage of the differential pair Q6-Q7 must have a positive temperature coefficient. This is achieved by making the transistor Q7 with an emitter area that is sufficiently larger than the emitter area of Q6. Moreover, it is clear that by adjusting the emitter area ratio of the transistors Q6 and Q7 and/or the ratio between R3 and R2, a stabilized voltage V2 may be obtained such that: δV2/δT=0. In the circuit diagram of Fig. 1, Q13, Q14, Q15, RE Q13 and R8 constitute a circuit that, through the local feedback, is capable of configurating substantially as a diode the transistor Q8, which, together with Q9, "reads" the differential stage Q6-Q7. The signal amplified by Q10 is transferred through the current mirror Qll and Q12 to the output node Vout and the resistances R7 and R6 close the general feedback loop, by feeding back the V2, voltage present on the intermediate node, to the base of Q7 of the input differential stage. EXAMPLE By assuming a Vbe Q2 =600mV, with a temperature coefficient of δVbe Q2 /δT=-2mV/°C, and a partition ratio given by R2/R1=0.1, a fractionary voltage is obtained that is given by: V1=Vbe Q2 R2/R1=60mV, having a thermal coefficient of: δV1/δT=-2mV/°C. By assuming n=2, Ae Q7 =10 and Ae Q6 =1, the following is obtained: ΔVbe = Vbe Q6 - Vbe Q7 ≈ 60mV and δΔVbe/ δT = +0.2mV/°C. Thefore, the circuit is capable of generating a stabilized voltage: V2=120mV, with a δV2/δT≈0. The voltage drop across R3A and R3B must be maintained equal to or lower than about 200mV, in order to ensure that the differential pair of transistors Q6-Q7 may function correctly without saturating. Of course, the circuit of the operational amplifier may be realized in a form different from the one depicted in the figures and described above. In particular, stages for correcting the "curvature" of the band-gap characteristic may be added, by employing a correction technique similar to the one described in the cited article: "A Curvature-Corrected Low-Voltage Band-gap Reference", IEEE Journal of Solid State Circuits, Vol. 28, No. 6, June 93, pages 667-670. The characteristic of a circuit made in accordance with the present invention is shown by the stabilized voltage V2 versus temperature curve of Fig. 3. In such an embodiment, without any correction stages, the output voltage V2 has a temperature coefficient that can be calculated as: δV2/δ°C= - 0.0833mV/°C

Claims: A circuit for generating a temperature stable reference voltage (Vout), summing a base-emitter (Vbe) voltage having a temperature coefficient of a first sign with a voltage proportional to the difference between two base-emitter voltages (ΔVbe), having temperature coefficients of opposite sign, for mutually canceling the effects of the two temperature coefficients of opposite sign and producing a sum voltage (Vout) without thermal drift, comprising an operational amplifier (Q6, Q7, I2, Q10, Q11, R5, Q12), configured as a noninverting buffer (R7, R6), means (Q8, Q13, Q14, Q15, I3, I4, RE Q13 , R3A, R3B, Q8) for controlling a bias current forced through a pair of input transistors (Q6, Q7) of said operational amplifier, characterized by further comprising a first circuit (Q0, Q1, Q3, RE Q1 , I1, R1, Q2, Q4, Q5, R2, RE Q5 ) producing a first voltage (V1) equivalent to a constant fraction that is less than unity of a base-emitter voltage (Vbe Q2 ); said non inverting buffer (Q6, Q7, 12, Q10, Q11, R5, Q12, R7, R6) producing on an output node a voltage (Vout) replica of the sum between said first fractionary voltage (V1) and a predefined and controlled intrinsic offset voltage of said pair of input transistors (Q6, Q7) which have an emitter area different from each other. The circuit as defined in claim 1, wherein said first circuit comprises a transistor (Q2) that generates a current proportional to its base-emitter junction voltage and a current mirror (Q2, Q5) capable of mirroring with a pre-established ratio a fraction of said proportional current on a resistance (R2), on which said first voltage (V2) is produced. The circuit as defined in claim 1, wherein said first voltage (V1) is applied to the base of a first transistor (Q6) of said differential input pair, having a control resistance (R3A) connected between a collector of said first transistor and ground; a fractionary voltage (V2) of the output voltage (Vout) of said operational amplifier, present on an intermediate node of a voltage divider (R7, R6) connected between the output node and ground, being applied to the base of the other transistor (Q7) of said differential input pair. The circuit as defined in claim 3, wherein said circuit for controlling a bias current forced through said input transistors (Q6, Q7) of different area comprises a current mirror (Q8, Q9, I3, I4, R3B, R3A) capable of forcing through said control resistance (R3A), connected between the collector of said first transistor and ground, a pre-established current (la). The circuit according to claim 4, wherein said current mirror comprises an amplifying stage (Q10, Q11), which drives an output stage (Q12) of the amplifier. A method for generating a reference voltage (Vout) summing a base-emitter (Vbe) voltage having a temperature coefficient of a first sign with a voltage proportional to the difference between two base-emitter voltages (ΔVbe), having temperature coefficients of opposite sign, for mutually canceling the effects of the two temperature coefficients of opposite sign and producing a sum voltage (Vout) without thermal drift, of a value lower than a base-emitter junction voltage (Vbe), characterized by summing a voltage replica of the difference between two dissimilar base-emitter junction voltages (ΔVbe) with a pre-established fraction (Vbe/K) less than unity of a base-emitter junction voltage. The method according to claim 6, wherein said voltage replica of the difference between two dissimilar base-emitter voltages is an intrinsic input offset voltage of an differential input pair of transistors (Q6, Q7) of a noninverting, buffer-configured, operational amplifier (Q6, Q7, I2, Q10, Q11, R5, Q12, R7, R6), the offset voltage of which is controlled by a local feedback loop (Q8, Q9, I3, I4, R3B, R3A) regulating the bias current that is forced through said input transistors (Q6, Q7).

Inventors: RICOTTI GIULIO, ROSSI DOMENICO, RICOTTI, GIULIO, ROSSI, DOMENICO
Applicants: ST MICROELECTRONICS SRL, STMICROELECTRONICS S.R.L.
Application year: 1993
Main classification: G05F3
IPC codes: G05F3, H03F1