Method and apparatus for theft detection using digital signal processing

Abstract: The present invention relates to a method and an apparatus for detecting the presence of a target (30) in an interrogation zone (24) utilizing digital signal processing. The signals received by an electronic surveillance system and whose frequency components have been phase shifted from a filtering operation (62) are restored by passing them into a signal delay circuit, tapping the delay circuit at several points therealong into associated signal channels, selectively amplifying or attenuating the signal in each channel and combining the signals in each channel.

Description: Field of the Invention This invention relates to the processing of electrical signals and in particular it concerns a method and an apparatus for utilizing digital signal processing in electronic theft detection. Description of the Prior Art United States Patent No. 4,623,677 to Pierre F. Buckens and assigned to the assignee of the present invention discloses and claims methods and apparatus for detecting the unauthorized taking of objects from a protected area, such as a store. Articles taken from the store must pass through an interrogation zone into which electromagnetic interrogation energy is continuously radiated. If, while an article is brought through the interrogation zone, it has an active target mounted thereon, the target will respond to the electromagnetic interrogation energy in the zone and will produce disturbances of that energy in the form of pulses having unique characteristics. These pulses are detected by a receiver at the interrogation zone. Subject - matter according to the preamble portions of present independent claims 1 and 8 is known from the said US-A-4 623 877. The apparatus disclosed in this document comprises a transmitter for generating an electromagnetic radiation in an interrogation zone, a receiver constructed and arranged to receive and detect the electromagnetic radiation which occurs in the interrogation zone from targets and other equipment, and to produce electrical signals corresponding to the detected radiation, a filter for filtering from the electrical signals selected frequency components, and a detector to detect the presence of electrical signals which correspond to the presence of a target in the interrogation zone. The filters used in this known apparatus are optimized for phase linearity to produce a phase shift or delay which is more linearly related to frequency so that the spreading in time of the sharp pulses produced by the targets in the interrogation zone is minimized. SUMMARY OF THE INVENTION The present invention provides additional improvements to the above-mentioned known apparatus. More specifically, the present invention makes target responses in an electronic article surveillance system more detectable by means of signal processing which substantially eliminates selected frequency components from energy to be detected and then replaces the original phase relationships among the remaining components, thereby preserving the unique characteristics of signals produced by the special targets attached to articles to be protected. According to the present invention there are provided a method and an apparatus for detecting the presence of a target in an interrogation zone. These method and apparatus comprise the steps of and apparatus for detecting all of the electromagnetic radiation which occurs in the interrogation zone being generated by a transmitter and disturbed by the presence of a target and/or other equipment, and producing electrical signals corresponding to the radiation, the electrical signals being made up of several frequency components; filtering from the electrical signals selected ones of the several frequency components; shifting the phases of the none-filtered frequency components in an adaptive equalizer by amounts such that the frequency components are returned to the same relative phase relationship which they had to each other prior to filtering; and detecting in non-filtered ones of said frequency components, the presence of electrical signals which correspond to the presence of a target in the interrogation zone. Fig. 1 is a perspective view of an electronic theft detection system embodying the present invention as installed in a supermarket; Fig. 2 is a diagrammatic view of the general components of the system of Fig. 1; Fig. 3 is a block diagram of the components of the system of Fig. 1; Fig. 4 is a series of waveforms showing the relative timing of signal processing in the system of Fig. 1; Fig. 5 is a further block diagram of a noise blanker portion of the system of Fig.4; Fig. 6 is a block diagram of long and short term averagers used in the system of Fig. 1; and Fig. 7 is a further block diagram of a pulse straightener portion of the system of Fig. 3. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT The present invention is applicable to any electronic article surveillance system in which a target causes rapid periodic electromagnetic disturbances. However, for purposes of illustration the invention will be described in conjunction with a so-called "magnetic" system in which an alternating magnetic field is introduced into an interrogation zone and targets on protected articles carried through the zone are driven alternately into and out of magnetic saturation by the alternating magnetic field. This produces periodic electromagnetic disturbances at frequencies which are harmonics of the original alternating magnetic field frequency. These harmonics, or selected ones of these harmonics, are detected and used to actuate an alarm. The arrangement shown in Fig. 1 is used in a supermarket to protect against theft of merchandise. As shown, there is provided a supermarket checkout counter 10 having a conveyor belt 12 which carries merchandise, such as items 14 to be purchased, past a cash register 16, as indicated by an arrow A. A patron (not shown) who has selected goods from various shelves or bins 17 in the supermarket, takes them from a shopping cart 18 and places them on the conveyor belt 12 at one end of the counter 10. A clerk 19, standing at the cash register 16, records the price of each item of merchandise as it moves past on the conveyor belt. The items are paid for and are bagged at the other end of the counter. The theft detection system according to this embodiment of the invention may include a pair of spaced apart antenna panels 20 and 22 next to the counter 10 beyond the cash register 16. The antenna panels 20 and 22 are spaced far enough apart to permit the store patron and the shopping cart to pass between them. The antenna panels 20 and 22 contain transmitter antennas which are simply loops or coils of wire or other conductive material capable of generating magnetic fields when electrical currents pass through them. These antennas generate an alternating magnetic field in an interrogation zone 24 between the panels. The antenna panels 20 and 22 also contain receiver antennas, which are also conductive coils capable of converting incident electromagnetic energy to electrical currents. These receiver antennas thus produce electrical signals corresponding to variations in the magnetic interrogation field in the zone 24. The antennas are electrically connected to transmitter and receiver circuits contained in a housing 26 arranged on or near the counter 10. There is also provided an alarm, such as a light 28, mounted on the counter 10, which can easily be seen by the clerk and which is activated by the electrical circuit when a protected item 14 is carried between the antenna panels 20 and 22. If desired, an audible alarm may be provided instead of, or in addition to, the light 28. Those of the items 14 which are to be protected against shoplifting are provided with targets 30. Each target 30 comprises a thin elongated strip of high permeability easily saturable magnetic material, such as permalloy. When protected items 14 are placed on the conveyor belt 12 they pass in front of the clerk 19 who may record their purchase. The items 14 which pass along the counter 10 do not enter the interrogation zone 24 and they may be taken from the store without sounding an alarm. However, any items which remain in the shopping cart 18, or which are carried by the patron cannot be taken from the store without passing between the antenna panels 20 and 22 and through the interrogation zone 24. When an item 14 having a target 30 mounted thereon enters the interrogation zone 24, it becomes exposed to the alternating magnetic interrogation field in the zone and becomes magnetized alternately in opposite directions and driven repetitively into and out of magnetic saturation. As a result, the target 30 disturbs the magnetic field in the interrogation zone in a manner such that pulses of magnetic energy are formed. These pulses, which are made up of frequency components at harmonics of the original or fundamental transmitted frequency, have a unique form, which makes it possible to detect their occurrence. The magnetic fields in the interrogation zone, including those which form the above described pulses, are intercepted by the receiver antenna which produces corresponding electrical signals. These electrical signals, as well as other internally generated electrical signals, are processed in the receiver circuits in a manner such that those produced by true targets can be distinguished from those produced by other electromagnetic disturbances and other internally generated electrical signals. Upon completion of such processing, the signals produced by true targets are then used to operate the alarm light 28. Thus the clerk 19 will be informed whenever a patron may attempt to carry unpurchased protected articles out of the store. Fig. 2 is a diagrammatic representation of the system of Fig. 1 as seen from a position along the path of movement through the interrogation zone 24. As indicated, transmitter circuits 40 are connected to a transmitter antenna 42 on one side of the interrogation zone 24; and a receiver antenna 44 on the other side of the zone 24 is connected to receiver circuits 46. These receiver circuits in turn are connected to an alarm 48. It has been found preferable to provide transmitter and receiver antennas on both sides of the zone 24; but for purposes of illustration and explanation Fig 2 shows a single transmitter antenna on one side and a single receiver antenna on the other side. The transmitter circuits 40 generate a continuous alternating electrical signal in the form of a sine wave and at a fixed fundamental frequency, for example, 218 HZ. This electrical signal is converted by the transmitter antenna 42 into a corresponding alternating magnetic interrogation field in the interrogation zone 24. The transmitted interrogation field is represented by the waveform I near the transmitter antenna 42. As can be seen, this waveform is in the shape of a sine wave. A target 30 in the interrogation zone 24 disturbs the field transmitted by the transmitter antenna and produces small pulses P as shown in a waveform II near the receiver antenna. The waveform II is basically the same shape as the waveform I except that the waveform II is slightly displaced in time due to its transit time across the interrogation zone. Further, the waveform II has pulses superimposed thereon which are caused by the target 30 in the zone. It should be noted that the waveform II, which has the same fundamental frequency as the waveform I, is synchronized with the wave form I. In addition, the pulses P in the wave form II are also synchronized with the waveform I. These pulses are actually the sum of several frequency components which are harmonics of the fundamental frequency of the transmitted magnetic field. The receiver antenna 44 converts magnetic fields which are incident thereon, including the waveform II, to corresponding electrical signals. These electrical signals are processed in the receiver circuits 46 to ascertain whether the magnetic field disturbances are those which have been caused by the presence of a true target 30 in the interrogation zone 24. If so, the receiver circuits send a signal to actuate the alarm circuit 48. It should be understood that in addition to the magnetic field from the target 30 which produces the waveform II, there are several other magnetic fields incident on the receiver antenna 44. These other fields may be caused by spurious electromagnetic disturbances from electrical equipment such as motors, lights, radio transmission, etc., or even by "innocent" objects, such as shopping carts or other metallic objects which disturb the magnetic field produced by the transmitter antenna 42. In addition, internally generated electrical disturbances alter the electrical signals produced by the receiver antenna 44. The system described herein uses various signal processing techniques to distinguish those disturbances produced by the presence of a true target 30 in the interrogation zone from the above mentioned other disturbances. Some of these techniques have been used in the past. The present invention makes it possible to process the received electromagnetic signals without significant phase or delay distortion due to filtering so as to maintain the characteristic shapes of the received signals. These features will become apparent from the following description of the internal configuration of the transmitter and receiver circuits. The overall block diagram of the transmitter and receiver circuits 40 and 46 is shown in Fig. 3. A clock generator 50 and a divider 52 are provided to synchronize the overall operation of the system. In this example the clock generator is chosen to produce pulses at a rate of 13,952 pulses per second on a sample clock signal line 51. The divider 52 is connected to the sample clock signal line 51 and is constructed to produce one output pulse for every 64 input pulses, that is, 218 pulses per second on a cycle clock signal line 53. The pulses from the divider 52 are applied to a low pass filter 54 which converts them to a continuous sine wave of 218 HZ. This sine wave is applied to an amplifier 56 which is connected to drive the transmitter antenna 42. The transmitter antenna 42 thus generates a continuous alternating magnetic field in the interrogation zone 24 as indicated by the waveform I in Fig. 2. The clock pulse generator 50, the divider 52, the low pass filter 54 and the amplifier 56 are all individually well known and no special form of any of these components is needed or desired in order to carry out the invention according to the best mode contemplated by the inventors. Electromagnetic energy from the interrogation zone 24, including disturbances produced by a target 30, if present, as well as other electromagnetic disturbances that may be present, are received by the receiver antenna 44 and converted to corresponding electrical signals. These signals are applied to front end amplifier and filter circuits 60. These front end circuits are designed to remove or reduce unwanted components from the electrical signals generated by the receiver antenna 44, particularly the very large fundamental frequency of the transmitter signal (i.e. 218 HZ). The front end circuits 60 are also individually well known and no special form is needed to carry out the invention. As mentioned, the front end amplifier and filter circuits 60 remove or reduce the very large fundamental frequency component, i.e. the 218 HZ component. For this purpose a notch filter has been found to be the simplest and most effective way to reduce this component. The front end amplifier and filter circuits 60 are connected through a first training/normal operation switch 61 (to be described more fully hereinafter) to internal amplifier and band-pass filter circuits 62. The purpose of these circuits is to attenuate frequency components above and below a predetermined frequency band. It has been found that those frequency components below the tenth and above the seventeenth harmonic of the fundamental frequency can be attenuated and the remaining components will closely represent the major distinctive features of the target produced pulses. Also, by attenuating the components above the seventeenth and below the tenth harmonic, a large portion of the interfering electrical energy from non-target sources is removed. The internal amplifier and band-pass filter circuits 62 are also well known and no special construction thereof is considered to be the best mode for carrying out this invention. In the illustrated embodiment the filter portion of the internal amplifier and band-pass filter circuits 62 is made up of a 9th order Butterworth highpass filter with a cutoff frequency of 2 KHZ (kilohertz) and a 9th order 0.01 db (decibel) Chebyshev lowpass filter with 3db down or -3db at 3800 HZ cutoff. The output of the internal amplifier and band-pass filter circuits 62 is connected to an analog to digital converter 64 which produces a digital output corresponding to the amplitude of the signal from the circuits 62 at any instant. The output from the analog to digital converter 64 is applied to each of M processors 65. Each processor comprises noise blanker circuits 67 and long and short term averager circuits 68. The output of each processor 65 is applied to a corresponding input 70a I ...70a M of a sample demultiplexer 70; and the single output of the sample demultiplexer 70 is applied to an adaptive equalizer 72. In the illustrative embodiment, which is presently preferred the number M is chosen to be sixty-four, which accommodates sixty-four samples during each cycle of the fundamental frequency. The amplifiers and filters 60 and 62 are designed to pass the 10th through 17th harmonics of the fundamental frequency and to attenuate frequency components above and below this band. Because of the characteristics of the filters, frequency components up to the 32nd harmonic may be passed to some appreciable degree. Therefore, to ensure against aliasing, the sampling and processing by the M processors 65 is at a rate substantially in excess of twice that frequency, namely, the 64th harmonic. The output of the adaptive equalizer 72 is applied through a full wave rectifier 73 to a signal channel 74, which contains a signal gate 76 and a low pass filter 78, and a noise channel 80, which contains a noise gate 82 and a peak detector 84. The outputs of the signal and noise channels 74 and 80 are compared in a comparator 86; and the comparator output is applied to the alarm 48. The signal and noise gates 76 and 82 are opened to pass signals along their respective signal and noise channels 74 and 80 at alternate times by gate signals from a gate generator circuit 88. The gate generator circuit 88 in turn receives pulses from the divider 52. The portion of the system following the adaptive equalizer 72, namely the portion containing the full wave rectifier 73 and the signal and noise channels 74 and 80 is, in principle, the same as described in the above referred to United States Patent No. 4,623,877 to Pierre F. Buckens, except that it is preferably implemented using well known digital circuits. Here it should be understood that while the processors 65, the sample demultiplexer 70, the adaptive equalizer 72 and the remaining components are all shown and described herein using block diagrams, the functions of these items in actual practice would be carried out by means of solid state integrated circuit components formed on chips that have been specially programmed to perform the functions to be described. It should also be understood the actual manner of programming the integrated circuit components is not part of the invention nor does it concern the best mode of carrying out the invention. Any programmer of ordinary skill in the art can program solid state components to perform the functions to be described; and there are many different ways of carrying out this programming, with no particular one being considered to be better than any other. The first training/normal operation switch 61 has a first input terminal 61a which is connected to the output of the front end amplifier and filter circuits 60, a second input terminal 61b which is connected to the output of a test pulse generator 63 and a common output terminal 61c which is connected to the input of the amplifier and bandpass circuits 62. The switch 61 is controlled by a programmed training/normal operation control unit 151, which also controls a second training/normal operation switch to be described hereinafter in connection with the adaptive equalizer 72. As shown, the adaptive equalizer 72 is also connected to receive signals from the training/normal operation switch control unit 151. Thus, depending on the setting of the first training/operation switch 61, signals are directed to the amplifier and bandpass filters 62 either from the receiver antenna 4 and front end circuits 60 or from the test pulse generator 63. The test pulse generator 63 is connected to receive cycle clock signals from the output of the divider 52 and to produce from each of these pulses a pulse similar to that which would come from the front end circuits when a true target 30 is present in the interrogation zone. During a "training" period, prior to normal operation of the system, the training/operation switch 61 is set with its second input terminal 61b connected to its common output terminal 61c and the pulse signals from the test pulse generator 63 are at this time applied to the amplifier and band pass circuits 62. During normal operation of the system, the switch 61 is set with its first input terminal 61a connected to the common output terminal 61c, so that signals from the receiver antenna 4 and the front end circuits 60 are applied to the amplifier and band pass circuits 62. Before describing the sample clock multiplexer 66, the noise blanker circuits 67, the averager circuits 68, the sample demultiplexer 70 and the adaptive equalizer 72, the general manner in which the system analyzes incoming signals will first be described in connection with Fig. 4. Waveform (a) of Fig. 4 represents the magnitude of the transmitted magnetic interrogation field which alternates at the fundamental frequency, which is the illustrative embodiment is 218 HZ. Waveform (b) of Fig. 4 represents the magnitude of an idealized signal incident on the receiver antenna 44 when a target 30 is present in the interrogation zone 24. As can be seen, the signal is dominated by the waveform of the alternating magnetic interrogation field from the transmitter antenna 42. This alternating magnetic field is at the transmitter or fundamental frequency of 218 HZ. The presence of the target 30 in the interrogation zone causes slight disturbances (P) of the magnetic field as a result of the target 30 being driven into and out of magnetic saturation twice during each cycle. A large portion of the signal produced by this alternating magnetic field at the fundamental frequency (218 HZ) is eliminated by the notch filter in the front end amplifier and filters 60. However, some remaining portion of this signal component is still present. The internal amplifier and band-pass filters 62 further attenuate the remaining portions of the fundamental frequency component as well as other components below the 10th harmonic and above the 17th harmonic of the fundamental frequency. Thus the output of the internal amplifier and band-pass filters 62 is made up of those frequency components which they pass, namely those components between 2,180 HZ and 3,706 HZ. While this is only a portion of the total spectrum of the frequency components of the pulses produced by the target 30, it has been found that this portion of the spectrum contains a sufficient amount of the components peculiar to the target 30. Accordingly the portion of the frequency spectrum between the 10th and the 17th harmonics of the fundamental frequency is well suited for accurate target discrimination. The waveform (c) of Fig. 4 is an idealized representation of true target pulses with the frequency components below the 10th and above the 17th harmonics removed. However, the actual form of the pulses is more like that shown in the waveform (d) of Fig. 4. This is because the filtering produced by the circuits 60 and 62 causes the retained frequency components to become phase shifted with respect to each other. Thus, the resulting pulses are spread out in time. According to the invention this pulse spreading effect is compensated so that several closely spaced pulses can be separately analyzed. In carrying out the present invention, the signals from the internal amplifier and bandpass circuits 62 are sampled at several instances during each transmitter cycle. It will be recognized that the more samples that are taken during each transmitter cycle, the closer the samples will follow the actual pulses resulting from the disturbances produced by the target 30. It has been found however that as long as the samples are taken at a rate which is greater than twice the frequency of the highest harmonic carried in the sample, the resulting sample composite will contain sufficient information to reproduce the pulses without any aliasing effects. In consideration of attenuation characteristics of the circuits 60 and 62, particularly the low pass filtering produced in the circuit 62, and in consideration of the resolution of the analog to digital converter 64 (e.g. twelve bits), a sampling rate of 64 times the fundamental frequency of 218 HZ is considered sufficient to avoid, for all practical purposes, the effects of aliasing. Thus the signals produced by the target 30 occur at a first frequency, namely, twice the fundamental frequency of the transmitter, which in this embodiment is 218 HZ. The frequency components which are used to ascertain the distinctive characteristics of the target signals extend up to a second, higher, frequency, which in this illustrative embodiment is the 17th harmonic, namely 3,706 HZ. The attenuation provided by the filters in the system effectively eliminate, or at least reduce to below an appreciable level, all frequency components below a third, still higher frequency, which in this illustrative embodiment, is the 32nd harmonic, namely 6,976 HZ. To avoid aliasing, samples are taken at a frequency of at least twice the third frequency, namely, the 64th harmonic or 13,952 HZ. As indicated in Fig. 3, there are provided as many noise blanker circuits 67 and signal averager circuits 68 as there are samples to be taken during each cycle; and each of these circuits is assigned to a corresponding sample interval. Thus, the sample clock multiplexer 66 has a single input terminal 66a at which the sample clock signal from the clock generator 50 is applied, and 64 outputs 66b l ...66b M each connected to a corresponding one of the noise blankers 67 and averager circuits 68. Thus the multiplexer 66 switches the clock signal on its common input terminal 66a to each of its output terminals 66b l ...66b M at a rate of 13,952 times per second or 64 time during each cycle of the fundamental interrogation frequency (218 HZ). Since an integral number (M) samples are taken during each cycle of the interrogation field and since the switching of the sample multiplexer 66 repeats after every M samples, and since each sample from the analog to digital converter 64 is made available to the noise blanker 67 in each of the M processors 65, each of the noise blankers 67 and signal averagers 68 operate on the sample associated with only an associated one of the M corresponding portions of successive magnetic field interrogation cycles. The circuitry of Fig. 3 eliminates signals which do not have a sufficient degree of consistency from cycle to cycle of the interrogation field. When a true target 30 passes through the interrogation zone 24 it produces pulses in corresponding portions of each interrogation field cycle. Since the interrogation field cycle is 218 -1 seconds (0.0046 seconds), a true target, whose passage time when carried through the interrogation zone is about 1.5 seconds, would ideally experience about 326 interrogation cycles and may produce about that many pulses. Actually, magnetic nulls are encountered along most paths so that less than 326 interrogation cycles are capable of producing target responses. It has been found that if only three pulses occur in a sequence of three successive interrogation cycles and if those pulses all have quite similar amplitude, it is likely that they were produced by a true target passing through the interrogation zone and not by a passing spurious electromagnetic disturbance or by some other energy source which is not synchronous with the magnetic interrogation field. However, a greater number of pulses from a correspondingly greater number of cycles may be compared to provide an even finer degree of selectivity. The processing of several signal samples from corresponding parts of several successive interrogation cycles to ascertain the presence of a true target is not new. According to the circuitry of Fig. 3, however, the successive samples are not processed in a manner which merely gives a weighted sum of those signals. Instead, the successive samples are compared in a manner which takes into account their deviation from each other. In other words, the consistency of sample amplitude from cycle to cycle is used as a criterion to ascertain whether the signals are being produced by an object which has been energized by the transmitter as opposed to one whose exitation originated from an outside source not associated with the system. When only an arithmetic average is used, a very large spike in one cycle may be sufficient to raise the signal level for several cycles by an amount to indicate the presence of a target, even though a target may not be present. However if the deviation from cycle to cycle is taken into account then the very large spike can be discounted. As specifically carried out, the circuitry of Fig. 3 processes the amplitudes of the samples taken at corresponding portions of N successive signal samples (for example, N=3 cycles), to ascertain whether the square of the sum of the sample amplitudes is greater than a predetermined constant K th (threshold constant), multiplied first by the same number of cycles, and multiplied further by the sum of the squares of the sample amplitudes. Typically, the constant K th has a value between 0 and 1 and may be supplied to the system in a manner which renders it field-adjustable. If the square of the sum of the sample amplitudes is greater, the system will allow the latest signal sample amplitude to pass through to the averagers for further processing, and at the same time will hold the value of the sample for comparison in the same manner with sample amplitudes which will be taken from corresponding portions of subsequent interrogation cycles. If the square of the sum of the sample amplitudes is less than the latter value, the system will not allow the sample amplitude to pass through to the averagers but it will hold the sample value for comparison in the same manner with sample amplitudes which will be taken from corresponding portions of subsequent interrogation cycles. Instead, it will feed back to the averagers the output of the long term averager for the selected sample interval. The noise blanker block diagram of Fig. 5 shows the construction of the noise blanker 67 which makes the above described comparisons-. As can be seen in Fig. 5, there is provided, for each of the noise blanker circuits 67, a summer 90 which, at one input terminal 90a, receives inputs from the analog to digital converter 64. The summer 90 also receives, at a second input terminal 90b, negative values of long term averager signals. The significance of these last mentioned long term averager signals will be described hereinafter. The summer 90 supplies its outputs to storage elements 94 1 , 94 2 , 94 3 (up to N such elements). Each element is activated by an output of the cycle clock multiplexer 92. The output of the sample clock multiplexer is connected to a common input terminal 92a of a cycle clock multiplexer 92. The cycle clock multiplexer 92 uses signals from the cycle clock signal line 53 to switch its sample clock multiplexer signal input terminal 92a to each of its output terminals 92b 1 ...92b N in succession, although, as mentioned above, sample amplitudes from only three successive cycles are taken in the present embodiment to obtain an indication as to whether any of them were produced by spurious or non synchronous energy. Therefore the cycle clock multiplexer 92 has three output terminals 92b 1 , 92b 2 and 92b 3 . For certain applications it may be desired to provide a finer resolution of the distinction between spurious or non synchronous energy and synchronous energy. In such case a larger number N of output terminals up to 92b N from the cycle clock multiplexer may be provided along with the associated additional elements shown connected by dashed lines. It should be understood that the cycle clock multiplexer 92, like the sample multiplexer 66, recycles, so that the next cycle clock transition to occur after the multiplexer has been switched to its last output terminal, causes the multiplexer to be switched again to its first output terminal. The output terminals 92b 1 ...92b N of the cycle clock multiplexer 92 are connected to associated signal storage devices 94 1 , 94 2 , 94 3 ...94 N . The storage devices are capable of holding the value of the sample last applied to their input terminal 94 1a , 94 2a , 94 3a ...94 na . This signal value appears continuously at the respective storage device's output terminal 94 1b , 94 2b , 94 3b , 94 nb . However, when the storage device's input terminals 94 1a , 94 2a , 94 3a ...94 Na become active, the old sample value in the storage device is replaced by the new value provided by the value at the summer output terminal 90c. The sample values in the signal storage devices are applied continuously to a sample value summer 100 where they are combined arithmetically. The resulting arithmetic sum is then applied to a squaring circuit 102 which produces an output corresponding to the square of its input. The squaring circuit 102 thus produces an output corresponding to the square of the sum of the successive sample values. The output of the squaring circuit 102 is applied to a plus input terminal 104a of a comparison circuit 104. The sample values in the signal storage devices 94 1 , 94 2 , 94 3 ...94 N are also applied to individual squaring circuits 106, 108, 110, etc. which, respectively, produce output values corresponding to the square of the values of the signals applied to their input. The outputs of the squaring circuits 106, 108, 110, etc. are applied continuously to a sample squared summer circuit 112 which produces an output value corresponding to the arithmetic sum of its inputs. The output of the sample squared summer 112 is thus a value corresponding to the sum of the squares of the values stored in the storage devices 94 1 , 94 2 , 94 3 ...94 N . The output of the sample squared summer 112 is applied to a multiplier circuit 114 where its value is multiplied by a number N, corresponding to the number of signal storage devices (in this embodiment, three), and by a preset value K th , which represents the threshold of signal value consistency needed to prevent a pulse from passing to the averagers. Typically, K th varies from 0 to 1. The output of the multiplier circuit 114 is applied to a negative input terminal 104b of the comparator circuit 104. The comparator circuit 104 is applied to a switch actuation terminal 116a of an inhibit switch 116. The inhibit switch 116 has a first signal input terminal 116b which is connected to receive the same signals which are applied from the analog to digital converter 64 to the input terminal 90a of the summer 90. The inhibit switch 116 also has a second signal input terminal 116c which is connected to receive signals from a long term averager to be described. When the output of the comparator circuit is more positive than negative, that is, when the square of the sums in the storage devices 94 1 , 94 2 , 94 3 ... 94 N is greater than the sum of the squares of those signals times N times K th , its output causes a common terminal 116d of the switch 116 to be connected to its first signal input terminal 116b so that the common terminal 116d receives signals directly from the analog to digital converter 64. However, when the output of the comparator circuit is more negative than positive, its output causes the common terminal 116d of the switch 116 to be connected to its second signal input terminal 116c so that its common terminal receives signals only from the long term averager (to be described). The signals from the analog to digital converter 64 which are applied to the noise blankers 67 are composite signals which include a first component of known periodicity, namely, the period separating alternate target produced responses, and a second component not of the known periodicity, namely, that resulting from other sources. The noise blankers compare the amplitudes of the composite signals from corresponding time intervals in each of a plurality of signal periods and operate their respective switches 116 to control the flow of the composite signals to further processing circuits, namely, the signal averagers 118 and 120, according to the degree of variation in those amplitudes. The components of known periodicity are closely similar to each other in amplitude from cycle to cycle; and if they predominate, the noise blanker will move the switch 116 to its upper position to pass the composite signal to the further processing circuits. If, however, the components which are not of the known periodicity predominate, they will not be similar in amplitude from cycle to cycle and the noise blanker will move the switch 166 to its lower position so that the composite signals will not pass to the averager circuits 118 and 120. The common terminal 116d of the switch 116 in the noise blanker circuit 67 is connected, as shown in Fig. 6, to both a short term averager 118 and a long term averager 120. The short term averager 118 includes a first multiplier 122, a summer 124, a delay register 126 and a second multiplier 128. The first multiplier 122 is connected to receive signals passed by the noise blanker circuit via the common switch terminal 116d and to multiply them by a preset value (1-A s ). The output of the first multiplier 122 is applied to the summer 124 which adds it to a value from the second multiplier 128. The sum of these values is applied to an input terminal 126a of the delay register 126 which stores them and maintains the summed value at an output terminal 126b until it receives a pulse from the sample clock multiplexer terminal 66b, which is dedicated to it. Because of the sample clock multiplexer logic, each output is activated for only one sample interval per cycle. Each averager is thus dedicated to a specific one of M sample intervals and is updated only during that one interval in each cycle. The output from the delay register 126 is applied to the second multiplier 128 where it is multiplied by a preset value (A S ). The multiplied value is then applied to the summer 124. In operation of the short term averager 118, signal values applied to the first multiplier 122 from the noise blanker circuit 67 are multiplied by (1-A S ) in the first multiplier 122, summed in the summer 124 with the output of the second multiplier 128, delayed in the delay register 126 and multiplied by the value (A S ) in the second multiplier.128. The output is then recycled through the summer 124, the delay register 126 and the second multiplier 128. This produces, at the output of the delay register 126, an output which is a weighted sum of the values of the previous input signals from the noise blanker circuit 67. The value of the each previous input signal diminishes in the short term averager 118 according to the number of times it circulates through the averager and according to the value of A S . If A were zero then each previous input signal would go to zero on its first recirculation and the value of the present input from the noise blanker circuit would be the new output. This is the shortest possible averaging. However, as the value of As increases, the previous input signal values have greater influence and the averaging period becomes longer. The long term averager 120 is of the same construction as the short term averager 118, and like the short term averager, the long term averager 120 comprises a first multiplier 130 which receives signals from the noise blanker circuit 67 and multiplies them by a preset value, which in this case is designated (1-A L ). The resulting value is added in a summer 132 with an output value from a second multiplier 134 and the summed value is applied to a delay register 136. The delayed output from the delay register 136 is multiplied by a preset value A L and applied to the summer 132. The only difference between the long and short term averagers 118 and 120 is the value of A. The value of A L in the long term averager 120 is greater than the value of A S in the short term averager 118 so that the long term averager takes into account a longer duration of past signal values in producing an output value. As mentioned above, the output from the sample clock multiplexer 66b, which is dedicated to this averager causes the output to be updated over every M sample interval. The output of the short term averager 118 is taken from the output of its delay register 126 and is applied to a plus input terminal 138a of an averager summing circuit 138. At the same time, the output of the long term averager 120 is taken from the output of its delay register 136 and is applied to a minus input terminal 138b of the averager summing circuit 138. The output of the averager summing circuit 138 is taken from an output terminal 138c and is applied to a corresponding input terminal 70a 1 ...70aM of the sample demultiplexer 70 (Fig. 3). The output of the long term averager 120 is also applied to the negative input terminal 90b of the summer 90 in the noise blanking circuit 67 (Fig. 5). As mentioned above, the noise blanking circuits 67 operate to prevent passage of any signals unless the values of at least three successive pulses applied thereto have a certain minimum variation. This will tend to block non-synchronous energy, that is energy which does not vary in synchronism with the transmitter. However, there are at times, other non target energy sources nearby which, for periods of three or more successive pulses, vary only minimally but which have a low average value over the period of the associated short term averager 118. That is, they do not persist as long as a signal from a target but while they do occur they may possibly not vary substantially from pulse to pulse. The signals produced by these energy sources are attenuated by both averagers 118 and 120. The difference of the outputs from the signal averagers 118 and 120 eliminates the effects of unvarying non-target synchronous energy sources, such as are produced by metal objects in the range of the transmitted magnetic fields or are produced internally by the circuit elements which operate synchronously with the transmitter. The average value of this unvarying energy is measured in each long term averager 120 and is subtracted from the output value of the corresponding short term averager 118 in the averager summing circuit 138. Since both averagers contain identical estimates of these unvarying energy sources, those signals are cancelled at the output of the differential summer 138. The outputs of the long term averagers 120, as mentioned above, are applied to the negative input terminal 90b of the summer 90 in their associated noise blanking circuits 67. The purpose for this is to keep the noise blanking circuits sensitive to variations in the pulse to pulse signal values. If the signal values of successive pulses vary by a given amount, that amount will be quite significant if the total signal value of each pulse is small. But if each pulse is added to the same large amount, for example from a non target energy source, then that same variation between the successive pulses will become relatively less significant. Therefore, by subtracting from the incoming pulses, the long term average value of the energy in the associated sample interval, the pulse to pulse variation is made more significant. The outputs from each of the averager summing circuits 138 are combined in the sample demultiplexer 70 (Fig. 3). Each of the averager summing circuit output terminals 138c are connected to a corresponding input terminal 70a 1 ...70aM of the demultiplexer 70. The demultiplexer 70 has a switch actuation terminal 70b connected to receive pulses from the sample clock signal line 51. These pulses cause the input terminals 70a 1 ... 70a M to be switched, in sequence, to a common output terminal 71. Thus the signals from the analog to digital converter, which were divided into time increments by pulses from the clock generator 50, and separately processed in the noise blankers and averagers, are reconstructed in the sample demultiplexer 70. By way of further explanation, in the transmitter portion of the system, the clock generator 50 produces a signal whose frequency is D*F 0 , where D is an integer and F 0 a frequency in hertz. This signal is divided by the dividers 52 to produce a signal of F 0 hertz. The F 0 hertz signal is then further processed, amplified and applied to the transmitter antenna 42 to create a field capable of exciting the target 30. The sole restriction on the method of processing F 0 is that the resulting transmitter field excites the target in such a manner as to produce a response which is periodic in F 0 . In the receiver, the receiver antenna 42, which is capable of sensing the presence of the target 30, is coupled through a series of filters and amplifiers which enhance the ratio of target signal energy to non-target signal energy. The accordingly enhanced output of these elements is presented to the analog to digital converter 64. The analog to digital converter generates sample signals at a rate of D*F 0 , where the D*F 0 signal is either obtained or derived from the system transmitter or independently generated in such a manner that the transmitter and receiver versions are identical in frequency. It should be noted that there are no restrictions on the phase relationship between these signals. The digital conversions of the analog to digital converter are presented to a functional block which includes a processor capable of performing digital signal processing functions at high speeds. The processor processes the signals applied to it in a manner which produces a condition representative of the presence of target, and activates the alarm 48 under that condition. The purpose of the noise blanking circuits is to distinguish between energy which is not a result of the transmitter's F 0 -based signal and which therefore is non system-synchronous, and that which is system-synchronous, with a view toward blocking the former from passing further in the signal processing chair. It does this by dividing the F 0 cycle into D time slots and making use of the fact that system-synchronous energy appears repeatedly in the same slot or slots, while non system-synchronous noise does not and is randomly spaced in time. It is important to distinguish between transient synchronous noise, such as that which occurs when targets or "innocent" objects are carried through the system, and stationary synchronous noise, which is always present. The latter is generally the result of spurious energy coupled from the transmitter to the receiver and of objects permanently mounted near the system's active region and responsive to the transmitter field. The following is a simplified description of the noise blanker algorithm in which the possible presence of stationary synchronous noise is ignored. The complete noise blanker algorithm, in which the presence of possible stationary synchronous noise is present, will be given later. In the system, N cycles of analog to digital conversions are stored in memory, there being D samples in every cycle. A sample in the d(th) slot of the n(th) cycle can be referred to as s nd . A software pointer advances through each cycle, one time slot at a time. When it reaches the Dth slot in a cycle, it advances to the next cycle. At the end of the Nth cycle, the pointer returns to the first slot of the first cycle. The pointer moves at a rate of D*F 0 , once for every analog to digital conversion. As the pointer moves to the next slot, the algorithm proceeds by computing the ratio of the square of the sum of all the samples of column d to N times the sum of the squares of the column d samples. Mathematically, this is written as: The value K can be seen to be a measure of how similar the sample values are within a column. The more similar, the higher the value of K, corresponding to a system-synchronous signal. It can be seen, for instance, that if all sample values within the current column are identical, then K = 1. If, however, the samples differ, and their average value is 0, then K = 0. By evaluating the above equation and determining whether K is greater than a given threshold K th , the algorithm determines whether the single sample being pointed to is synchronous, and therefore should be passed on for further examination, or non-synchronous, whereby it is deemed noise and unworthy of further processing. In practice, it is simpler to avoid division and evaluate the computationally equivalent problem: The above would be sufficient if it were not for the existence of stationary synchronous energy in real systems. This energy manifests itself by adding to each sample a component of energy which does not change with cycle n, but rather is constant within a column d. This background energy necessitates the modification of the above equations. In order to properly account for this term, it is necessary to first develop an estimate of it. Such an estimate may be obtained through the use of a synchronous filter or averager. A synchronous filter (synchronous with D*F 0 , that is) can be developed by dividing the F 0 cycle up into D time slots, there being a one to one correspondence between each averager slot and each column of slots developed in the simplified noise blanker algorithm. As the sample pointer detailed above advances from slot to slot, a separate pointer to the averager advances with it in lockstep. However, when the simplified noise blanker algorithm pointer advances to the first sample of the next cycle, the averager pointer merely returns to the first sample of the averager. Before detailing how the averager works in conjunction with the noise blanker algorithm, operation of the averager as a stand alone device will be described. Each output sample a d of a stand alone averager is combined with an input x d and is modified according to the following equation: III. a d ⇐ a d ×α + x d ×(1 - α) where alpha is a constant between 0 and 1 which establishes the time constant of the filter. The averager thus acts to produce for each time slot an average of the energy incident upon each of its D cells. It should be noted here that the averager input x d is in fact the output of a modified version of the noise blanker algorithm which takes into account the averager output state. The following set of equations describes the output y d of the full noise blanker algorithm for the arbitrary time where all pointers are in column d: VI . M d - K th ×V d If the above difference is positive, then: VII . y d ⇐ s nd VIII . a d ⇐ a d ×α + x d ×(1-α) If the difference is negative, then: IX . y d ⇐ a d and X . a d ⇐ a d The signals from the common output terminal 71 of the demultiplexer 70 are applied to the adaptive equalizer 72 which is shown in more detail in Fig. 7. Here again it should be understood that while the adaptive equalizer is shown in block diagram in Fig. 7, this is for purposes of illustration; and the actual device is formed as part of an integrated circuit. As shown in Fig. 7, the adaptive equalizer 72 includes a delay line register 140 which receives signals at an input terminal 140a from the output terminal 71 of the sample demultiplexer 70. The delay line register 140 has a series of cells 140b 1 ...140b M ; and the signals applied at the input terminal 140a at one end of the register 140 pass through each of the cells in step by step sequence as clock pulses are applied from the sample clock signal line 51 (Fig. 3) to a clock pulse terminal 140c. The delay line register 140 should have a total length or delay period equal to the period of the fundamental frequency, namely the frequency of the interrogation magnetic field; and the number of cells 140b should be equal to the number of pulses M applied to the terminal 140c during such period. Thus the delay line register 140 contains, at any instant, the signal pulses which have passed through the noise blankers and averagers during one cycle of magnetic interrogation field variation. Each cell in the delay line register 140 has a tap output 140x 1 ...140X M which is connected to an associated output multiplier 142 1 ...142 M . These multipliers 142 accept as inputs, signals from associated tap coefficient lines 141 1 ...141 M . Those signals are generated by the M amplitude control adjustment circuits 154 1 ... 154 M only one of which, 154 1 is shown. The outputs of the multipliers 142 1 ...142 M are combined in a summing circuit 144. The summing circuit 144 has a common output terminal 144a which is connected to a common terminal 146a of a second training/operation switch 146. One output terminal 146b of the training/operation switch 146 is connected to the full wave rectifier 73 (Fig. 3). Another output terminal 146c of the training/operation switch 146 is connected to a plus input terminal of a summing circuit 150. An idealized pulse signal D M , from an internal source (not shown) is applied to a negative terminal of the summing circuit 150. It has been found that a delta function which consists of a signal with a single non-zero value in one of M sample intervals and a value of zero elsewhere is not itself a useable signal for this application. For a delta function to be useful, frequency components which have already been filtered out by the filters 62 would have had to be present. Instead, it has been found that a useful signal may be obtained by sampling a signal of the shape shown in Fig. 4c. In the present embodiment, nine of the M samples (M=64) in this sequence are non-zero and correspond to the pulse shown. This produces a significant improvement in the shape of the pulse over that which exits from the filters 62, as shown in Fig. 4d. When the second training/operation switch 146 is in the train position (that is, when the common terminal 146a is connected to the second output terminal 146c), the summing circuit 150 subtracts the value of the idealized pulse signal from the value of the signal in the summing circuit 144. The resulting signal, which represents an error value, is applied to a multiplier 152, which multiplies it with a coefficient 2W. By choosing a large value for W it becomes possible to achieve rapid convergence or adaptation of the adaptive equalizer 72. However, the precision of adjustment is low in such case.. On the other hand, by choosing a small value for W, the precision of adjustment is increased but the speed at which it occurs is reduced. It is beneficial to provide a value of W which varies with the amount by which the adaptive equalizer deviates from the ideal setting. Then, for large deviations, the adjustments will be large and rapid, and as the amount of deviation decreases, the resulting value is applied to each of several individual amplitude control adjustment circuits 154 1 ...154 M associated with each of the cells in the delay line register 140. For purposes of clarity of explanation only one of the amplitude control adjustment circuits 154 1 is described in connection with Fig 7. However, the construction and operation of the others is the same. As shown in Fig. 7, the amplitude control adjustment circuits 154 each comprise a multiplier 156, an adder 158 and a delay register 160. The multiplier 156 is connected to receive and multiply the value of the output from the multiplier 152 with the value of the output signal 140x from an associated delay register cell 140b. The resulting value is added in the adder 158 to the tap coefficient 141 which was developed during the time of the preceding input from the clock pulse generator 50. The output from the adder 158 is supplied to the storage register 160 where it is delayed for a duration equal to one sample interval, namely, the pulse period of the sample clock signal line 51. The output of the storage register is the tap coefficient 141 and is applied to the associated multiplier 142. As mentioned above, when the second training/operation switch 146 is switched to its operation position, namely with the common terminal 146a connected to the second output terminal 146b, the output signals from the adaptive equalizer are supplied through a full wave rectifier to the signal and noise channels 74 and 80. These signals can pass through the respective channels only at alternate times and only when the signal and noise channel gates 76 and 82 are opened. These gates are opened by outputs from the gate generator 88 which in turn receives pulses from the divider 52 (Fig. 3). The gate generator 88 is set so that it opens the signal gate 76 during that portion of the magnetic interrogation wave cycle within which pulses from true targets are likely to occur, that is, when the magnetic field is close to being changed in direction and is at relatively low intensity. The gate generator 88 opens the noise gate 82 when the magnetic interrogation field is in the portions of its cycle where it has a high intensity, namely, an intensity beyond that at which a true target would produce pulses. The signals which pass through the signal gate 76 are applied to the low pass filter 78 which provides smoothing. The smoothed signals are then applied to the plus input terminal of the comparator 86. Meanwhile the signals which pass through the noise gate 80 are applied to the peak detector 84 which produces an output along the noise channel 80 corresponding to the value of the signal which occurred while the noise gate 82 was last opened. This noise signal value is applied to the minus terminal of the comparator 86. The comparator 86 will produce an alarm output when the value of the filtered signal in the signal channel 74 is greater than the value of the signal in the noise channel 80. The alarm output is then applied to actuate the alarm 48. Operation of the above described system occurs in two modes, namely, a training mode and an operation mode. The purpose of the training mode is to preset the amplitude control adjustment circuits 154 and the signals on the associated tap coefficient lines 141 1 ...141 M in the adaptive equalizer 72. This training mode occurs for a period of about 15 seconds when the system is first turned on. During this time the training/normal operation control unit 151 switches the first and second training/operation switches 61 and 146 to their training position, which allows the storage elements 160 to be updated at each sample interval. That is, the first switch 61 is set to connect the output of the test pulse generator 63 to the amplifier and bandpass filters 62 (Fig. 3) and the second switch 146 is set to connect the output of the adaptive equalizer summing circuit 144 to the summing circuit 150 (Fig. 7). After this training has been concluded the unit 151 returns the movable element of the switch 61 (Fig. 3) to the input terminal 61a and the movable element of the switch 146 (Fig. 7) to its output terminal 146b. It also sends a signal to the storage registers 160 to prevent them from being further updated; and the registers hold their present value. The purpose for the training mode is to set the adjustable tap coefficients in the adaptive equalizer 72 so that the adaptive equalizer will compensate for the phase distortion that occurs during the passage of signals through the amplifier and bandpass filters 62. As mentioned previously, these circuits remove frequency components outside a frequency range which is used to ascertain the distinctive characteristics of target produced pulses. This enables the pulses to be sampled and processed digitally; provided however, that they are sampled at a frequency at least twice the highest frequency passed by the amplifier and bandpass filters 62. In filtering out the high and low frequency components however, the filters also shift the relative phases of the signal components that they do pass. The adaptive equalizer 72, when its tap coefficients are properly set, compensates for this phase shifting. The setting of these adjustable amplitude control devices is carried out during the training mode, namely for the first fifteen or so seconds after the system is turned on and while the first training/operation switch 61 is set to connect the output of the test pulse generator 63 to the amplifier and bandpass filters 62 and while the second training/operation switch 146 is set to connect the output of the summing circuit 144 in the adaptive equalizer 72 to the summing circuit 150 and the following amplitude control adjustment circuits 154 and while the storage registers 160 are being updated in each sample interval. The adaptive equalizer 72 operates in the manner of a finite response (FIR) or transversal filter having a tapped delay line with taps that are variously weighted and summed to produce an output. The setting of these taps is accomplished by interactively adjusting them according to a stochastic gradient algorithm to correct signals supplied from the test pulse generator 63 and bring them into conformity with a stored idealized pulse D M with minimal phase distortion. The idealized pulse D M is supplied from a pulse generator (not shown) and applied to the negative input terminal of the summing circuit 150 (Fig. 7) where it is algebraically combined with the output of the summing circuit 144 to generate an error signal. The error signal is scaled in the multiplier 152 and then supplied to each of the amplifier control adjustment circuits 154. Each amplifier adjustment control circuit multiplies the value of the modified error signal with the value of the signal from its associated tap output 140 X and, in the adder 158, adds the result to the tap coefficient value 141 obtained during the last sample interval. The output of the adder 158 is then stored in the storage register 160 for one sample period, namely, the pulse period of the clock generator 50, for use in the next operation. Meanwhile, the result from the previous sample, which is at the output of the storage register 160, is applied to the associated multiplier 142 and adjusts its amplification or attenuation by a predetermined increment. By repetitively sampling, comparing and adjusting, as above described for a period of several seconds, the several multipliers 142 are set to compensate for the effects of phase shifting produced by the amplifier and bandpass filter circuits 62. The tap coefficients then remain at their respective settings thereafter while the system is switched to its normal mode of operation by changing the setting of the first and second training/operation switches 61 and 146 to their respective normal operation settings and precluding the storage registers 160 from further modification. The switches 61 and 146 may be operated by the preprogrammed control circuit 151 shown in Fig. 3. It should be understood that the general idea of use of a delay line or delay register with multiple taps and adjustable tap coefficients to reshape a pulse signal is known. However, the adaptation of such general technique to the detection of signals from targets in electronic article surveillance is believed to be novel. There has thus been described a novel system for detecting the presence of targets in an interrogation zone and in the presence of non-target produced electrical and electromagnetic energy. According to the invention the system automatically compensates for the effects of filtering on the phase relationships of different frequency components of the portions of the signals being analyzed in the system. It should be understood however, that the noise blanker circuits 67, both alone and in combination with either or both the long term and the short term averager, and the adaptive equalizer circuit 72, with its automatic adjustment feature could be used in other applications.

Claims: A method of detecting the presence of a target (30) in an interrogation zone (24), said method comprising the steps of: (a) detecting all of the electromagnetic radiation which occurs in said interrogation zone (24) being generated by a transmitter (40, 42) and disturbed by the presence of a target (30) and/or other equipment, and producing electrical signals corresponding to said radiation, said electrical signals being made up of several frequency components; (b) filtering from said electrical signals selected ones of said several frequency components; and (c) detecting, in non-filtered ones of said frequency components, the presence of electrical signals which correspond to the presence of a target (30) in said interrogation zone; characterised in that before step (c), the phases of said non-filtered frequency components are shifted in an adaptive equalizer (72) by amounts such that said frequency components are returned to the same relative phase relationship which they had to each other prior to filtering. A method according to claim 1, wherein said non-filtered frequency components have, at successive times, corresponding magnitudes, and wherein said step of shifting said non-filtered frequency components comprises altering said corresponding magnitudes by predetermined amounts and combining said altered magnitudes. A method according to claim 2, wherein said step of altering said corresponding magnitudes comprises directing said non-filtered frequency components through a delay circuit (140) having taps therealong, recovering a signal sample at each of said taps simultaneously, selectively altering the magnitude of each signal sample and combining said altered signal samples. A method according to claim 3, wherein said step of altering comprises the step of passing said signals through multipliers. A method according to claim 3, wherein said step of altering the magnitude of each signal sample comprises passing each signal sample through a signal multiplier whose other input is a tap coefficient. A method according to any of claims 3 to 5, wherein said step of combining said altered signal samples comprises summing the magnitudes of said altered signal samples. A method according to claim 1, wherein said step of shifting said non-filtered frequency components includes, in a training mode, the further steps of: applying electrical test signals, which are ideally representative of a target (30), to said adaptive equalizer (72), comparing the output of said adaptive equalizer (72) to a signal representative of a proper output to produce an error signal, and adjusting said adaptive equalizer (72) to minimize said error signal. An apparatus for detecting the presence of a target (30) in an interrogation zone (24), said apparatus comprising: a transmitter (40, 42) for generating an electromagnetic radiation in said interrogation zone (24); a receiver (44, 46) constructed and arranged to receive and detect all of the electromagnetic radiation which occurs in said interrogation zone (24) being generated by said transmitter (40, 42) and disturbed by the presence of a target (30) and/or other equipment, and to produce electrical signals corresponding to said radiation, said electrical signals being made up of several frequency components; a filter (62) connected to filter from said electrical signals selected ones of said several frequency components; and a detector to detect, in non-filtered ones of said frequency components, the presence of electrical signals which correspond to the presence of a target (30) in said interrogation zone (24), characterised in that said apparatus further comprises an adaptive equalizer (72) constructed and arranged to shift the phases of said non-filtered frequency components by amounts such that said frequency components are returned to the same relative phase relationship which they had to each other prior to filtering. An apparatus according to claim 8, wherein said adaptive equalizer (72) is connected to receive and detect the magnitudes of said non-filtered frequency components which occur at successive times, to alter said detected magnitudes by predetermined amounts and to combine said altered magnitudes. An apparatus according to claim 9, wherein said adaptive equalizer (72) comprises a delay circuit (140) having taps therealong to recover signal samples from different locations, simultaneously, along said delay line, and signal altering elements (142 1 -142 M ) connected to said taps to selectively amplify or attenuate the magnitude of the signals passing therethrough. An apparatus according to claim 10, wherein said signal altering elements (142 1 -142 M ) are multipliers. An apparatus according to claim 10, wherein said adaptive equalizer (72) includes a signal summer (144) connected to sum the magnitudes of said altered signal samples. An apparatus according to any of claims 8 to 12, wherein said apparatus further includes a signal generator (63) for generating idealized pulse signals representative of signals produced by an ideal target (30) in said interrogation zone (24) and a training/operation switch (61) connected to supply signals to said filter (62) alternately from said signal generator (63) and from said receiver (44).

Inventors: LUNDQUIST DAVID T, PAUL CHRISTOPHER R, LUNDQUIST DAVID, T., PAUL CHRISTOPHER, R.
Applicants: SENSORMATIC ELECTRONICS CORP, SENSORMATIC ELECTRONICS CORPORATION
Application year: 1993
Main classification: G08B13
IPC codes: G08B13