Transconductor stage

Abstract: A transconductor stage for high-frequency filters operated on a low voltage supply, being of a type which comprises an input circuit portion having signal inputs, further comprises a pair of interconnected differential cells (2,3) being associated each with a corresponding signal input. Each cell incorporates at least one pair of bipolar transistors (Q1,Q2;Q3,Q4) having at least one corresponding terminal, e.g. the emitter terminal, in common.

Description: This invention relates to an improved transconductor stage for high frequency filters operated on a low voltage supply, being of a type which comprises first and second differential cells each comprising a first and a second transistor having a common terminal, each first transistor having a respective control terminal receiving an input signal, said second transistors having respective control terminals connected together, the first transistor of the first cell and the second transistor of the second cell having respective terminals connected together to provide a first current output, the first transistor of the second cell and the second transistor of the first cell having respective terminals connected together to provide a second current output. The field of application of this invention is particularly, but not solely, related to monolithically integrated time-continued filters, and the description to follow will make reference to that field of application for convenience of illustration. Discussion of the Prior Art As it is known, a number of techniques for providing time-continued integrated filters have been proposed. Among them, those which appear at present to perform best in high frequency environments utilize a transconductor stage as the filter basic block. Most likely, this is attributable to the voltage-to-current conversion being carried out in an open loop within the transconductor, and to the so-called non-dominant pole of the converter not being limited by the frequency unitary gain of the amplification. In any case, the performance features which are usually expected of a transconductor stage, for high-frequency applications, are basically the following: maximum amplitude of the signal to be processed; low noise generation; low biasing consumption; high transconductance gm value; high dynamic range of transconductance gm; and high gain, in excess of 50 dB. The prior art has already provided a transconductor stage design which has good performance features closely approximating the above target features. Reference is directed in particular to a transconductor described in Italian Patent Application No. MI91A 003018 by this Applicant. That Application discloses a transconductance amplifier provided in a mixed bipolar/MOS technology and comprising an input circuit portion having a differential cell with a pair of source-sharing MOS transistors which are connected to an output circuit portion comprising a pair of base-sharing bipolar transistors. While being in many ways advantageous and substantially achieving its objective, this prior solution has a drawback in that high transconductance values cannot be obtained concurrently with a low current draw and good linearity. In addition, this prior solution is unsuitable for operation at low power supply values, e.g. of no more than 3 Volts. A demand definitely exists at present for analog circuits which can operate on just 3 Volts and yet afford comparable performance levels to circuits operated on 5 Volts. Some solutions have been proposed in the prior art which would seemingly operate even on supply voltages close to 3 Volts; however, under such operating conditions, these solutions have proved ineffective to ensure control of the transconductance gm value over a sufficiently wide range. An improved transconductor structure is disclosed in the UK patent application No. GB 2 195 211 which relates to an amplifier substantially as defined in the preamble of the present application. A similar structure is disclosed also in the U.S. patent No. 4,052,679 which relates to a phase shifting circuit for shifting the phase of an alternating circuit. Both the above prior art references provide a pair of differential amplification circuits, each having a pair of transistors with a common terminal. However, both solutions don't provide a transconductor stage with high linearity when supplied with voltages less than 3 Volts. The underlying technical problem of this invention is, therefore, that of providing a transconductor stage which has such structural and functional features as to overcome the above-noted drawbacks of prior art solutions. Summary of the Invention This technical problem is solved by a transconductor stage of the type indicated hereinabove and defined in the characterizing parts of the appended claims. The features and advantages of a transconductor stage according to the invention can be more clearly understood from the following detailed description of an embodiment thereof, given by way of example and not of limitation with reference to the accompanying drawings. Brief Description of the Drawings In the drawings: Figure 1 is diagramatic representation of a transconductor stage according to the invention; Figure 2 is a diagramatic representation of a second embodiment of the transconductor stage in Figure 1; Figure 3 is a further diagramatic view showing in greater detail the transconductor in Figure 2; Figure 4 is a diagramatic view of the transconductor stage of this invention; and Figures 5 and 6 are diagramatic views of respective examples of time-continued filters incorporating the transconductor stage of this invention. Detailed Description With reference to the drawing views, shown in diagramatic form at 1 is a transconductor stage embodying this invention for processing high-frequency, higher than 50 MHz, analog signals while operating on a low voltage supply Vd. The stage 1, which has been depicted in a simplified form, is preferably provided in the bipolar technology. It would be equally possible, however, to provide it in a mixed technology, that is by having both bipolar and MOS transistors incorporated. The stage 1 includes signal inputs, indicated at B and C, being each associated with a respective differential cell 2, 3. The inputs B and C are applied corresponding voltage signals Vi+ and Vi-; the differential voltage Vin at the stage 1 input is given by the difference (Vi+ - Vi-) between said input signals The first cell 2 comprises a first pair of bipolar transistors Q1, Q2 of the npn type having their respective emitter terminals E1, E2 in common to form a circuit node at which a constant bias current I is made available by means of a generator A1. The second cell 3 comprises a second pair of bipolar transistors Q3, Q4 of the npn type having their respective emitter terminals E3, E4 in common to form a second circuit node at which a constant current I is made available by means of a generator A2. The transistors Q1 and Q4 have equal minimum emitter areas. The transistors Q2 and Q3, while being the same size, have a larger area than the former transistors, Q1 and Q4. Specifically, the area of transistors Q2, Q3 is α times that of transistors Q1, Q4, and the value of α lies preferably between four and six. The cells 2 and 3 are connected to each other through their respective transistors. More particularly, the transistors Q2 and Q3 have their respective bases B2 and B3 in common and connected to ground via a constant voltage Vcm generator, such as a battery E. The voltage Vcm is called common mode voltage and intended for maintaining the bases B2 and B3 at a common bias value. The collector C1 of the first transistor Q1 in the first pair of transistors Q1, Q2 and the collector C4 of the second transistor in the second pair of transistors Q3 and Q4 are the outputs U1, U2 of the stage 1 wherethrough differential currents I01 and I02 are flowed. The collectors C2, C3 of the second transistor Q2 in the first pair and the first transistor Q3 in the second pair are connected to the second output U2 and the first output U1, respectively. The two cells 2 and 3 are therefore connected to each other in a substantially crossover fashion. The operation of the transconductor stage 1 according to the invention will now be described. An exemplary application thereof will be discussed later herein. The stage 1 biasing within the linear operation region of the transistors Q1,...Q4, to enable proper voltage-to-current linear conversion is ensured by suitably selecting the value for the bias current I. Using the well-known Ebers-Moll's equations, the operation of the inventive circuit can be described mathematically. For large signals, the transconductance gm value can be defined by the following relationship: gm = (I/V T ) [α * e x (α 2 +1)e 2x +4αe x +(α 2 +1) (α+e x )2(αe x +1) 2 ] where V T is the voltage equivalent of the absolute temperature (V T = 25 mV at 27°C), I is the bias current, α is the ratio of the emitter area of the transistors Q2 and Q3 with respect to the emitter area of the transistors Q1 and Q4, and x = (Vin/V T )*1/2. From equation (1), the transconductance gmo value for small signals, i.e. with Vin<<1, can be obtained as follows: gm o = (I/V T )*[2α/(α+1) 2 ] This equation (2) only holds, in fact, for small differential signals, but does bring out the dependence of the value of gm o on the bias current I. By changing the value of that current, the transconductance gmo value can be adjusted within a fairly wide range, e.g. of 1 to 10. The operational linearity of the stage 1 is instead adjusted by selecting the α parameter (the area of transistors Q2 and Q3). Actual tests and simulations conducted on the basis of equation (1) show that, for a value of α equal to 5.2, the signal harmonic skew can be kept quite small and close to 1% with input signals having an amplitude of 300 mV. This skewing remains small throughout the range of practicable bias current I values. Additionally, the aforesaid value of the α parameter has proved to be non-critical since, for variations in the 5 to 5.4 range, the skewing remains at all times close to 1%, when Vin=300mV. The circuit of this invention is useful in high-frequency environments on account of its simple structure and because it incorporates bipolar transistors. Understandably, it matters to the proper operation of the stage 1 that the building technology can provide npn bipolar transistors having a low base resistance, high transition frequency, and low time constant as given by the product of the collector resistance by the collector/substrate capacitance. These constructional features ensure good speed for the circuit 1. A second embodiment of the circuit shown in Figure 1 will now be discussed with specific reference to Figure 2, in which co-operating parts and details having similar structure and operation to the previously discussed embodiment are denoted by the same reference numerals. In the diagram of Figure 2, a first resistor R1 is connected between the respective bases B1 and B2 of transistors Q1 and Q2 in the first cell 2, while a second resistor R2 is connected between the respective bases B3 and B4 of transistors Q3 and Q4 in the second cell 3. Connected between said resistors R1, R2 and ground is a capacitor Cx. The connection point between the resistors and the capacitor Cx is marked as node A. Preferably, the resistors R1, R2 are the same value of 5 kΩ, and that of the capacitor Cx is 5 pF. According to the diagram of Figure 2, the bases B1 and B4 of the input transistors Q1, Q4 are forward biased by the signal from a preceding stage, that is a stage connected ahead of stage 1 in the direction of signal propagation. Further, it is assumed that the bias voltage provided by the stage ahead is equivalent to the common mode voltage Vcm that was provided by the battery E in the previous embodiment. On the other hand, the base voltage of the remaining transistors Q2 and Q3 is obtained from the voltage drop across the resistors R1 and R2. Assuming the current Ib which flows from the node A to the bases B2, B3 to be null, then the node A will attain a voltage exactly equal to Vcm, thereby duly biasing the stage 1. The capacitor Cx will, moreover, function as a filter for any high-frequency signals present at the node A. Actually, the current Ib drawn by the transistors Q2, Q3 will be definitely other than zero, and be a function of the collector current of said transistors, as well as of the temperature and the variables of the technological process. In essence, the current Ib generates a voltage offset between the inputs B, C and the node A; accordingly, this current will have a second-order effect on the linearity of the transconductor stage. The voltage offset is represented by the voltage drop VBA, being identical with the voltage drop VCA and equal to R*Ib/2. In order to obviate any problem from the presence of the voltage offset, the solution shown in Figure 3 is provided which may be regarded as an option for the stage 1. A bipolar transistor Q5 of the npn type is connected between the voltage supply Vd and ground, with the collector C5 being connected to the voltage supply and the emitter E5 connected to ground via a current generator I2. This transistor Q5 is selected to have the same area as the transistors Q2 and Q3; in addition, the current I2 on the collector C5 is equal to the collector current of those same transistors Q2 and Q3. The base B5 of the transistor Q5 draws a current which is Ib/2. Advantageously, a current mirror 6 is connected between the base B5 of the transistor Q5 and the node A. In a preferred embodiment, the current mirror 6 comprises a pair of P-channel MOS transistors. The current mirror 6 is balanced to provide a multiplication ratio of two. Thus, the node A will be supplied, through the mirror circuit 6, a current equal to Ib/2 multiplied by two, and hence identical with Ib. This fully compensates for the voltage offset. Therefore, with this solution as illustrated by Figure 3, the current at the resistors R1, R2 will disappear and the offset phenomenon be suppressed. An exemplary application of the transconductor stage according to the invention will now be described. In Figure 4, the stage 4 is depicted in a highly schematized form that only purports to illustrate it generally. Figure 5, using the basic diagram of Figure 4, shows an example of a time-continued low-pass filter 5 incorporating a number of transconductor differential stages embodying this invention and being denoted by the acronyms GM1, GM2, GM3, GM4. The filter 5 may be integrated monolithically and is specifically conceived for very high-frequency, above 50 MHz, applications. This filter 5 is a second-order filter, it comprising a so-called biquadratic cell. The stages 1 are cascade connected together with the outputs U1, U2 of one stage being connected to the respective inputs B, C of the stage next after. Advantageously, the outputs of the fourth stage are feedback connected, and in a crossover fashion to the inputs of the second stage, while the outputs of the third stage are feedback connected to the inputs of the third stage itself. Capacitive couplings are also provided between the outputs of the first two stages and ground. The filter 5 shown is specially effective where the dynamic range of the input signal is less than 300 mV peak-to-peak. With higher dynamic ranges, the use of buffer circuits 4 has shown to effectively avoid signal skewing. Shown in Figure 6 is an exemplary application wherein unitary gain buffers 4 have been incorporated to the filter 5, ahead of the signal inputs of the second and third filter stages. The transconductor stage of this invention does solve the technical problem and affords a number of advantages, foremost among which is, of course, its ability to process large signals in a linear fashion. The solution provided by the invention also allows of the transconductance gm value to be changed within a sufficiently wide range while operating with a voltage supply of 3 Volts only. The transconductance gm value is of about 2 mA/V with an overall bias current of 400 µA. The transconductor of this invention is characterized by having a parasitic pole at higher frequencies than 2 GHz with a gain close to 50 dB. Understandably, this solution may be altered and modified in several ways within the invention scope as set forth in the appended claims.

Claims: A transconductor stage for high-frequency filters operated on a low voltage supply, being of a type which comprises first and second differential cells (2, 3) each comprising a first (Q1, Q4) and a second transistor (Q2, Q3) having a common terminal (E1, E3), each first transistor (Q1, Q4) having a respective control terminal (B, C) receiving an input signal, said second transistors (Q2, Q3) having respective control terminals connected together, the first transistor (Q1) of the first cell and the second transistor (Q3) of the second cell (3) having respective terminals (C1, C3) connected together to provide a first current output (U1), the first transistor (Q4) of the second cell and the second transistor (Q2) of the first cell (2) having respective terminals (C4, C2) connected together to provide a second current output (U2) , characterized in that it comprises a respective current generator (A1, A2) connected to provide a fixed current to each common terminal, and that the first transistors (Q1, Q4) have equal minimum emitter areas while the second transistors (Q2, Q3) have emitter areas α times larger than the corresponding first transistors. A transconductor stage according to claim 1, wherein said α value lies preferably between four and six. A transconductor stage according to Claim 1, characterized in that said transistors (Q1,Q2;Q3,Q4) are npn bipolar transistors and said common terminal is the emitter terminal (E1, E3). A transconductor stage according to Claim 1, characterized in that, connected between the respective control terminals (B1,B2) of first and second transistors (Q1,Q2) in the first cell (2), is a first resistor (R1), and that connected between the respective control terminal (B3,B4) of first and second transistors (Q3,Q4) in the second cell (3), is a second resistor (R2). A transconductor stage according to Claim 4, characterized in that said resistors (R1,R2) have the same value. A transconductor stage according to Claim 4, characterized in that a capacitor (Cx) is connected between said resistors (R1,R2) and ground. A transconductor stage according to Claim 4, characterized in that it includes a current mirror (6) connected between said resistors and the control terminal (B5) of a further transistor (Q5) connected between a voltage supply (Vd) and ground. A transconductor stage according to Claim 7, characterized in that said current mirror (6) is formed of MOS transistors. A transconductor stage according to Claim 1, characterized in that said transistors are of the MOS type. A second-order low-pass filter (5), characterized in that it includes at least one differential transconductor stage (1) as claimed in Claim 1.

Inventors: ALINI ROBERTO, GORNATI SILVANO, MOLONEY DAVID, ZUFFADA MAURIZIO, ALINI, ROBERTO, GORNATI, SILVANO, MOLONEY, DAVID, ZUFFADA, MAURIZIO
Applicants: SGS THOMSON MICROELECTRONICS, SGS-THOMSON MICROELECTRONICS S.P.A.
Application year: 1992
Main classification: H03F3
IPC codes: H03F1, H03F3, H03H11