DC-TO-DC CONVERTER

Abstract: A high frequency switching converter has a primary mosfet switch (14) controlled by an oscillator (34), and operating in resonant reset mode. The secondary includes a magnetic amplifier (38) associated with a circuit (40, 42) which applies a clamped bias voltage to the magnetic amplifier (38) to limit its flux swing. The magnetic amplifier may have an auxiliary winding (38b) arranged to produce a pedestal voltage in the secondary circuit for further reduction of flux swing. The converter also has an overload protection arrangement (60-68) operating entirely on the primary side by reference to the rise time of the reset capacitor (Cr).

Description: "DC-to-DC Converter" This invention relates to dc-to-dc converters, particularly converters suitable for large-scale production at low cost. The invention is particularly directed at low power applications, but may also be applied to higher power applications. Low cost, low power dc-to-dc converters are commonly used as regulated power supplies in computer circuits and the like, and comprise basically a switching circuit, a transformer and an output filter. Our published European Patent Application EP 0,481,682A describes converters operating in resonant reset mode at about 1MHz and using an integrated magnetic structure which may be in the form of a simple balun. One object of the present invention is to improve on that disclosure by providing an excellent degree of voltage regulation at low cost, and by producing very low power dissipation. The use of magnetic amplifiers (magamps) as control elements in switch mode converters has previously been considered, but these have not been extensively used as they have required special quality ferrites and have been poor in terms of size and energy dissipation. Embodiments of the invention will now be described, by way of example, with reference to the drawings, in which:- Fig. 1 is a circuit diagram of a prior art converter using a magamp; Fig. 2 is a B-H curve illustrating the operation of the circuit of Fig. 1; Fig. 3 is a circuit diagram of a first embodiment of the present invention; Fig. 4 is a B-H curve illustrating the operation of the circuit of Fig.3; Figs, 5A and 5B are output voltage waveforms illustrating the operation of the circuit of Fig. 3; Figs. 6A and 6B are plan and side views, respectively, of a ferrite toroid for use in the circuit of Fig. 3; Fig. 7 is a circuit diagram of an alternative embodiment; and Fig. 8 is a circuit diagram of another embodiment. Fig. 1 shows a prior art converter. Input d-c voltage on lines 10, 12 is switched by MOSFET 14 to apply square waves to a transformer 16. The transformer 16 is reset by a resonant circuit comprising the transformer 16, the MOSFET 14 and a reset capacitor Cr. The output of transformer 16 provides a d.c. output at terminals 20, 22 via a low-pass filter formed by L2, Cl and C2. Regulation is provided by a magamp arrangement comprising a ' agamp 18 and a reset circuit 24. With this circuit, when the primary MOSFET 14 turns off, the secondary voltage of the transformer 16 switches to -Vsec. The current in the core of the 1 magamp 18 will then ramp down towards zero and if no 2 reset were present would actually reach zero with the 3 magamp flux moving to the residual flux point Br (see 4 Fig. 2) . When reset is applied to the magamp core from 5 the reset circuit 24 via reset diode 26, the voltage 6 across the magamp core will be Vsec - Vbias, with the 7 result that during the remainder of the off time the 8 magamp core will be reverse biased to a negative 9 magnetizing current and its flux density will be reset 10 to a value Be lower than Br. When the MOSFET 14 turns 11 on again, the magamp core is not in saturation and, 11 depending on how far it has been reverse biased, the 1, core * ill block a certain quantity of Volt-seconds, :.<-- thus ...lowing control of the output voltage. 15 16 It will be seen that, in this system, the flux 17 excursion is large from Bs to Br and then to Be with 18 control occurring between Br and Be. This leads to a 19 high degree of dissipation and in practice requires the 20 use of expensive special-purpose ferrites having a 21 square-loop B-H characteristic, to avoid the need for 22 large reset currents. 23 24 Turning to the embodiment of the present invention 25 shown in Fig. 3, the basic structure of the circuit is 26 similar to that disclosed in our above-referenced EP 27 0,481,682A, the disclosure of which is incorporated 28 herein by reference. Briefly stated, a single 29 integrated xneqnetic structure, suitably in the form of 30 a balun, is -sed to provide the coupling transformer 31 16, the outp:•-; low-pass filter choke L2, and primary 32 and secondary inductors 30, 32. The main switching 33 MOSFET 14 is controlled by a 4049 i.e. 34 operating as 34 an oscillator, preferably at 1 MHz, via an emitter- 35 coupled switching stage 36 (which may be omitted if a suitable MOSFET is used, such as a 2VN4210A) . The present invention differs from the circuits disclosed in EP 0,481,682A in that the secondary uses a magamp 38 to control regulation. The magamp 38 has a main winding 38a which is operated in set mode; it is clamped to the bias voltage Vbias applied across it by transistor 40 and series diode 42. The voltage across the magamp 38 no longer has the large excursion of the total secondary voltage swing Vsec-Vbias, and the magamp 38 is effectively clamped to a maximum voltage determined solely by the bias voltage. In this case, the total Volt-seconds across the magamp is Vbias x Ton, as opposed to (Vsec-Vbias) x Ton for the reset case. This gives a dramatic reduction in flux swing, and the effective control of the output occurs over the range Br - Bs. As seen in Fig. 4, by using an ordinary ferrite material with a non-square B- H loop, the use of the Br-Bs range for control is very effective. Returning to Fig. 3, the bias voltage on the magamp 38 is controlled by the signal applied to the base of transistor 40, which is derived by operational amplifier 42 from a reference input from Zener diode 44 and an output voltage signal from resistor chain Rl, R2. The circuit as thus far described gives a useful improvement over the prior art, enabling a very small, low cost ferrite to be used as a magamp core. Preferably, however, the circuit includes additionally the feature which will now be described. The magamp 38 has an auxiliary winding 38b connected to provide a pedestal voltage in the output waveform. The auxiliary winding 38b is returned to the switched input to the low-pass filter via diode 46. The result is illustrated in Fig. 5, in which Fig. 5A shows the voltage input to the low-pass filter at a 5W load, and Fig. 5B the same input at a 0.5W load. It will be seen that in each case there is a pedestal voltage 50 resulting from the operation of the auxiliary winding 38B, and a superimposed pulse 52 resulting from the operation of the main winding 38a, the length of the pulse 52 varying according to load. The length of the superimposed pulse 52 can vary from zero to 100% of the pedestal on time. The use of the pedestal voltage feature reduces the flux swing still further while retaining full control of the output voltage. The ratio of pedestal height to total height in the voltage waveform is a function of the turns ratio of the main and auxiliary windings of the magamp 38. In the embodiment shown the turns ratio is 1:1 which gives total height: pedestal height of 2:1. The maximum control range will also be 2 : 1 which is more than adequate for most applications. For optimum overall performance it is also preferred to use feedforward on the primary: this extends the total control range to nearly 3:1 and at the same time reduces the flux density in the main isolation transformer. This feedforward is very simply obtained by a series Zener diode 54 and resistor 56 connected between the input bus 10 and the timing input of the oscillator 34. The pedestal voltage arrangement has a further advantage in that it simplifies start-up. If only a singe magamp winding were used, it would be necessary to provide a separate starting arrangement (which might simply be a shunt resistor) . The core of the magamp 38 in this circuit can be of surprisingly small size, and of extremely low cost owing to the small size and non-square-loop material. In the foregoing embodiment, the windings 38a and 38b are each ten turns wound on a ferrite toroid. Fig. 6 shows the toroid 38c, the dimensions of which in this embodiment are A = 4 mm, B = 2 mm, C + 2 mm. This embodiment was designed to handle 5W, but can handle up to 10W, the power dissipation is only some mW, and at nominal input voltage, 5W load and 1 MHz switching frequency the temperature rise of the toroid 38c was 14°C. The foregoing system is very cost effective, simple to make, and has a regulation performance of 0.5% to 1.0% over any combination of 2:1 input voltage range and 10:1 power range. It is also very temperature tolerant: -changes of only a few mV in output voltage have been measured over the wide range of -40°C to +70°C. A further advantage of the foregoing circuit is that it has an excellent transient response. In the embodiment shown, the transient response to a step change in load. is of the order . of 20 microseconds, whereas prior art circuits have a transient response between 100 microseconds and 1 millisecond. The basis for this good transient response is the fact that the transient response is controlled only by the loop gain of the secondary magamp circuitry which, by its very simple nature and high speed capability, does not have any excessive unwanted phase shifting components inherent in its design; components of this nature would normally be detrimental to good transient response. The transient response is thus also helped by the fact that the control is entirely on the secondary side. A further aspect of the present invention is concerned with protection against short circuits and overload. Referring now to Fig. 3, a resistor 60 is connected in series with the resonating capacitor Cr. The voltage across resistor 60, which is a function of load current, is sampled and peak rectified by a Zener diode 62 and diode 64 in series. The rectified signal is filtered by a capacitor 66 and resistor 68, and the resultant signal is applied to a spare gate input of i.e. 34, the gate output being applied to the ring-of- three oscillator section of the i.e. 34. The gate threshold on a CMOS device is relatively tightly controlled and is typically 40% of the supply voltage. In operation, the current limit point is set by the parallel resistor 60 such that no current limiting occurs until approximately 20% overload. Beyond this, the d.c. voltage applied to the control gate progressively limits the on time of the system to reduce the output voltage and current to safe levels, even under short circuit conditions. This arrangement makes use of the fact that, with resonant reset switching and a high switching frequency, the rise time of the leading edge of the MOSFET drain voltage is a function of the output current. Effectively, the circuit described acts to differentiate the leading edge to produce a voltage pulse having an amplitude proportional to load current. Other means of thresholding and foldback may be used to produce the same result. This protection system is very simple and efficient, and well suited to low cost production. It also has the great advantage of being entirely on the primary side of the converter and thus avoiding signal transfer across the isolation boundary provided by the isolation transformer. Fig. 7 shows a further embodiment providing two output voltages. This is similar to the foregoing embodiment and like parts are denoted by like references. A second output channel 70 is provided, having a secondary winding 72 wound on the common balun. A magamp 74 for this channel is wound on a separate toroid similar to that of the magamp 38. The magamp 74 has no auxiliary winding; start-up is achieved via the main channel. A pedestal voltage is not used in channel 70 in the circuit shown, but could be incorporated by use of an auxiliary magamp winding. Fig. 8 shows another variation on the circuit of Fig. 3. The operational amplifier 42 is replaced by transistor 80 and Zener diode 82. This somewhat degrades voltage and temperature characteristics, but reduces cost and allows surface-mount devices to be used throughout. Fig. 8A shows a modification which improves on the voltage and temperature stability of Fig.8, the simple Zener diode 82 being replaced by a TL 431 Zener device 82a to give a more accurate breakdown voltage, used with an oppositely-poled diode 84 to cancel temperature effects of the Vbe of transistor 80. • - Modifications may be made within the scope of the invention as defined in the Claims. For example, it would be possible to fabricate the magamp as part of the integrated magnetic structure of the converter.

Claims: CLAIMS 1. A dc-to-dc converter comprising an isolation transformer having a primary and a secondary, switching means for switching an input voltage across the transformer primary, and an output circuit across the transformer secondary which includes a low-pass filter; the converter having regulating means comprising a magnetic amplifier connected between the transformer secondary and the low-pass filter, and means for applying a clamped bias voltage to the magnetic amplifier to limit the flux excursion in the magnetic amplifier. 2. A converter according to Claim 1, in which the magnetic amplifier has an auxiliary winding connected to the input of the low-pass filter to produce a pedestal voltage during current flow in the secondary. 3. A converter according to Claim 1 or Claim 2, in which the means for applying the clamped bias voltage comprises a transistor having its main path in series with a diode and responsive to the converter output voltage to vary the bias voltage. 4. A converter according to Claim 3, in which the transistor is controlled by an operational amplifier connected to said output voltage and to a voltage reference. 5. A converter according to any preceding Claim, including a second transformer secondary which supplies a second output circuit, the second output circuit including a second magnetic amplifier and means for applying a clamped bias voltage to the second magnetic amplifier. 6. A converter according to any preceding Claim, in which the magnetic amplifier has a core in the form of a toroid of a ferrite material having a non-square- loop B-H characteristic. 7. A dc-to-dc converter comprising an isolation transformer having a primary and a secondary, switching means for switching an input voltage across the transformer primary, and an output circuit across the transformer secondary which includes a low-pass filter; the switching means operating in resonant reset mode and including a switching device, an oscillator driving the switching device, and a reset capacitor connected in parallel with the switching device; the converter having a protection circuit comprising means for generating a signal representative of the rise time of the voltage on the reset capacitor, and threshold means responsive to said signal to control operation of the oscillator when said signal exceeds a predetermined threshold. 8. A converter according to Claim 7, in which the switching means is a MOSFET device operating at 0.5 - 1.5 MHz and the means for generating said signal comprises a resistor in series with the reset capacitor, the threshold means comprises a Zener diode sampling the voltage on the resistor, and the Zener diode output is differentiated and applied to a gate circuit controlling the oscillator.

Inventors: ATTWOOD BRIAN ERNEST, ATTWOOD, BRIAN, ERNEST
Applicants: VALOR ELECTRONICS INC, VALOR ELECTRONICS INC.
Application year: 1992
Main classification: H02M3
IPC codes: G05F1, H02M3