Cycloconverter and the method of controlling the same

Abstract: A method of controlling a cycloconverter, for supplying an AC power having a given frequency to a converter group (23 to 26) having at least two cascade-connected converters (23, 24 or 25, 26) and controlling a firing angle of each of the converters (23 to 26), thereby converting the AC power supplied to the converter group (23 to 26) into an AC power having a predetermined frequency, includes the steps of controlling an output voltage from a converter (24, 26) in one stage of the converter group (23 to 26) to obtain an AC output voltage including a synthetic voltage of an AC voltage having an amplitude equal to a difference between a maximum output voltage of the converter (23) and a DC voltage and a predetermined DC voltage in accordance with the polarity of a reference voltage signal having a predetermined frequency, controlling an output voltage from a converter (24, 26) in the other stage of the converter (24, 26) to obtain an AC output voltage obtained by subtracting the output from of the converter (24, 26) in one stage from the reference voltage signal, and continuously varying a voltage value of the DC voltage in accordance with the frequency of the reference voltage signal.

Description: The present invention relates to a cycloconverter for converting an AC power having a given frequency into an AC power having an arbitrary frequency and the method of controlling the same. A two-stage-cascade cyclic-current cycloconverter for driving an induction motor will be described below. FIG. 6 shows an arrangement of a main circuit of this cycloconverter. A main circuit 9 of the cycloconverter has U-, V-, and W-phase converter units in correspondence with U, V, and W phases of an induction motor 13, respectively. In the U-phase converter unit, positive converters 23 and 24 for flowing a positive component of an AC output current are two-stage-cascade-connected to negative converters 25 and 26 for flowing a negative component of the output AC current, and transformers 21 and 22 are connected to the converters 24 and 26 in the first stage and the converters 23 and 25 in the second stage, respectively. The U-phase converter unit further includes a reactor 27 for suppressing a cyclic current. The V-phase converter unit is constituted by connecting transformers 31 and 32, positive converters 33 and 34, negative converters 35 and 36, and a reactor 37 in the same manner as in the U-phase converter unit. The W-phase converter unit is constituted by connecting transformers 41 and 42, positive converters 43 and 44, negative-converters 45 and 46, and a reactor 47 in the same manner as in the U-phase converter unit. Outputs from the U-, V-, and W-phase converter units of the main circuit 9 are connected to the induction motor 13. In the cycloconverter having the above arrangement, firing signals generated by an asymmetrical controller 20 are supplied to the positive and negative converters of each of the U-, V-, and W-phase converter units. For example, firing signals U1TA and U2TB for determining an output voltage U1S are supplied to the converters 24 and 26, respectively, in the first stage of the U-phase converter unit, and firing signals U2TA and U2TB for determining an output voltage U2S are supplied to the converters 23 and 25, respectively, in its second stage. The two converters of each of the positive and negative groups output equal voltages at the same time upon application of the firing signals, and an average voltage V of the voltage outputs from the two converters is supplied to the induction motor 13. This average voltage V is given by the following equation: V = (U1S + U2S) = (1/2)(U1SA + U2SB) + (1/2)(U2SA + U2SB) Although effective values of the output voltages from the positive and negative converters are equal to each other, a cyclic current flows from the positive converters 23 and 24 to the negative converters 25 and 26 due to a differential voltage caused by output voltage waveforms. The level of the cyclic current is suppressed by the reactor 27 for cyclic current suppression. The foregoing goes for the V- and W-phase converter units. The asymmetrical controller 20 will be described below. FIG. 7 shows waveforms of an output voltage E u 1S from the positive and negative converters in the first stage, an output voltage E u 2S from the positive and negative converters in the second stage, and a total output voltage E u of the U-phase converter unit. FIG. 8 is a flow chart for generating an output voltage reference. In order to obtain the output voltages E u 1S and E u 2S in the corresponding stages of the U-phase converter unit, the two converters in each stage are controlled as follows. That is, the output voltage of the two converters in one stage is fixed at a maximum voltage Eα of the converter, and the output voltage of the two converters in the other stage is so controlled as to obtain the total output voltage E u of the U-phase converter unit. Converter output voltage references in the respective stages are obtained by the following equations. That is, if the total output voltage reference of the converter unit is E u > 0: 1st-stage converter output voltage reference E1 u = Eα 2nd-stage converter output voltage reference E2 u = Esinϑov - Eα for |Esinϑov| ≦ 2Eα, where Eα is the maximum output voltage of the converter. If the total output voltage reference of the converter unit is E u ≦ 0: E u 1S = Esinϑov + Eα E u 2S = -Eα By controlling the output voltages from the converters as described above, the voltage to be applied to the motor acquires a sine wave as indicated by the waveform E u shown in FIG. 7. When the asymmetrical control as described above is performed, the converters in the first stage constantly generate the maximum voltage with a high input power factor. As compared with symmetrical control in which the same sine-wave output voltage is given to both the first and second stages, therefore, the input power factor can be largely improved. The symmetrical control system can be applied with no problem when a frequency of the output voltage reference is low, i.e., in a low-speed operation range. If, however, a frequency is as high as 20 to 30 Hz or more, for example, an output voltage waveform in each stage for outputting the above asymmetrical control cannot be correctly output. As a result, no sine wave can be obtained as an output voltage to a motor, and a distortion in sine wave becomes a disturbance such as a torque ripple. The above asymmetrical control, therefore, can be applied in only a low-frequency range. In order to realize a cycloconverter capable of driving a motor at a high speed and having a high input power factor, control must be performed such that the asymmetrical control is used to improve an input power factor as a cycloconverter in a low-speed range (at about a base speed) in which the input power factor is particularly degraded and the symmetrical control is used in a high-speed range to correctly supply an output waveform during a high-frequency operation. From prior art documents DE-A-2 543 066 and 1986 IEEE IAS CONFERECE RECORD, 28 September 1986, DENVER, pages 36-41, SAITO ET AL.: "A multi-processor-based, fully digital, AC drive system for rolling mills", it is known to use cycloconverters with cascade-connected converters which are controlled from different reference voltages to get different output voltages which are added to give a resulting output voltage. It is an object of the present invention to provide a cycloconverter and the method of the same capable of continuously, smoothly performing switching from asymmetrical to symmetrical control or vice versa in accordance with an operation range of a motor, thereby realizing a high-speed operation without disturbing an output voltage waveform, and having a high input power factor in a low-speed range. To solve this object the present invention provides a cycloconverter and a method of controlling a cycloconverter as specified in claims 1 and 10, respectively. Thus, the method of controlling a cycloconverter, for supplying an AC power having a given frequency to a converter group having at least two cascade-connected converters and controlling a firing angle of each of the converters, thereby converting the AC power supplied to the converter group into an AC power having a predetermined frequency, comprises steps of controlling an output voltage from a converter in one stage of the converter group to obtain an AC output voltage including a synthetic voltage of an AC voltage having an amplitude equal to a difference between a maximum output voltage of the converter and a DC voltage and a predetermined DC voltage in accordance with the polarity of a reference voltage signal having a predetermined frequency, controlling an output voltage from a converter in the other stage of the converter group to obtain an AC output voltage obtained by subtracting the output from of the converter in one stage from the reference voltage signal, and continuously varying a voltage value of the DC voltage in accordance with the frequency of the reference voltage signal. According to the present invention as described above, since switching from asymmetrical to symmetrical control can be continuously, smoothly performed, an input power factor can be improved in a low-speed operation range, and a high-speed operation with good output characteristics can be realized without disturbing an output voltage waveform in a high-speed operation range. This invention can be more fully understood from the following detailed description when taken in conjunction with the accompanying drawings, in which: FIG. 1 is a functional block diagram for driving an induction motor by using a cycloconverter according to the present invention; FIG. 2 is a flow chart for realizing a method of controlling the cycloconverter according to the present invention; FIG. 3 is a graph showing a relationship between a speed and a DC bias Vc for explaining an operation of the present invention; FIG. 4A is a timing chart for explaining an operation of the cycloconverter according to the present invention, showing first and second output voltage signals obtained in asymmetrical control; FIG. 4B is a timing chart for explaining an operation of the cycloconverter according to the present invention, showing the first and second output voltage signals obtained while asymmetrical control transits to symmetrical control or vice versa; FIG. 4C is a timing chart for explaining an operation of the cycloconverter according to the present invention, showing the first and second output voltage signals obtained in symmetrical control; FIG. 4D is a timing chart showing a reference voltage signal of the cycloconverter according to the present invention; FIG. 5 is a functional block diagram showing a control unit of the cycloconverter according to the present invention; FIG. 6 is a main circuit diagram of a cycloconverter; FIG. 7 is a timing chart for explaining an operation of a conventional cycloconverter; and FIG. 8 is a flow chart for realizing asymmetrical control of the conventional cycloconverter. FIG. 1 shows a circuit arrangement for driving an induction motor by a cycloconverter according to an embodiment of the present invention. This cycloconverter is constituted by a reference voltage generator for generating a reference voltage signal determined in accordance with the frequency of an AC power to be subjected to frequency conversion, a controller 8 for converting the reference voltage signal into a firing signal on the basis of a flow chart shown in FIG. 2, and a main circuit 9 having a plurality of converters each having a firing angle controlled by the firing signal supplied from the controller 8. The arrangement of the main circuit 9 is the same as that shown in FIG. 6. As will be described below, the reference voltage generator generates U-, V-, and W-phase reference voltage signals E u , E v , and E w having phases shifted through 120° from each other and outputs the generated signals to the controller 8. The controller 8 has functional blocks shown in FIG. 5 and operates on the basis of the flow chart shown in FIG. 2. The operation of the controller 8 will be described below by taking a U phase as an example. When the reference voltage signal E u having an AC waveform shown in FIG. 4 is input, the controller 8 starts processing shown in step S1 of FIG. 2. In step S1, a polarity determining section 51 determines a current polarity of the input signal. If the reference voltage signal E u falls within the range of the positive polarity, the flow advances to step S2, and a first output voltage generating section 52 generates first and second voltage signals E u 1S and E u 2S. The signals E u 1S and E u 2S are obtained by the following equations: E u 1S = (Eα - Vc)sinϑov + Vc E u 2S = E u - E u 1S where Eα is the maximum output voltage of the converter, Vc is the DC voltage component, ϑov is the voltage phase of the reference voltage signal E u . If the polarity of the reference voltage signal Eu falls within the range of the negative polarity, the flow advances to step S3, and a second output voltage generating section 53 generates the first and second voltage signals E u 1S and E u 2S. In this case, the signals E u 1S and E u 2S are calculated as follows: E u 1S = E u - E u 2S E u 2S = (Eα - Vc)sinϑov - Vc Note that the DC voltage component used in the arithmetic operation performed in steps S2 and S3 is varied by a DC component varying section 54 as a function of the speed of an induction motor 13. For example, as shown in FIG. 3, the DC voltage component Vc is set at the maximum output voltage Eα when the speed is set from 0 to a low-frequency operation speed ωα, continuously decreased from Eα to 0 when the speed is set from the low-frequency operation speed ωα to a high-frequency operation speed ωβ, and set at 0 when the speed is set at the high-frequency operation speed ωβ or higher. Although ωα and ωβ can be arbitrarily set within the range of 0 < ωα < ωβ, the frequency ωα is set within a frequency range where asymmetrical control can be performed without any problem for an output voltage distortion of the cycloconverter, and the high-speed operation frequency ωβ is set within a frequency range applicable to the system, wherein the output voltage waveform is distorted unless symmetrical control is executed. The speed of the induction motor 13 is in proportion to the frequency of the reference voltage signal E u . That is, if the speed is increased, the frequency of the reference voltage signal is increased, and vice versa. When the first and second voltage signals E u 1S and E u 2S are generated in step S2 or S3, the flow advances to step S4. In step S4, The signals E u 1S and E u 2S are converted into firing signals corresponding to their voltage levels by a table 55. That is, when the first voltage signal E u 1S is input, the table 55 outputs firing signals U1TA and U1TB which allow a synthetic output voltage (1/2)(U1SA + U1SB) of two converters in the first stage to be the first voltage signal E u 1S. Similarly, when the second voltage signal E u 2S is input, the table 55 outputs firing signals U2TA and U2TB which allow a synthetic output voltage (1/2)(U2SA + U2SB) of two converters in the second stage to be the second voltage signal E u 2S. The signals U1TA and U1TB are supplied to the two converters in the first stage. The signals U2TA and U2TB are supplied to the two converters in the first stage. The arrangement of the reference signal generator is shown in detail in FIG. 1. In the reference generator, as shown in FIG. 1, a deviation between an externally supplied speed reference ωr* and a real speed ωr of the induction motor 13 detected by a speed detector 12 is input to a speed controller 1. The speed controller 1 executes a PID arithmetic operation for the input deviation to obtain a torque reference T and outputs the obtained reference T to a divider 2. When the input real speed ωr of the induction motor 13 reaches a predetermined value, a flux arithmetic operation unit 3 calculates a flux reference  for decreasing the speed in accordance with an inverse proportional function. The divider 2 divides the torque reference T* by the flux reference  and supplies the quotient to a vector controller 4. On the basis of the output from the divider 2 and the flux reference , the vector controller 4 decomposes the primary current of the induction motor 13 into a torque current component I 1q * and a flux current component I 1d * by using the constants (a secondary resistance R2, a secondary inductance L2, and a mutual inductance M) of the induction motor 13. That is, the controller 4 multiplies the output from the divider 2 by L₂/R₂M to generate the torque current component I 1q *. In addition, the controller 4 divides the flux reference  by the mutual inductance M and supplies the quotient to a flux saturation function circuit 4a. In accordance with a relationship between the flux reference  of the induction motor 13 and an excitation current Io, the circuit 4a calculates the excitation current Io in consideration of saturation of the magnetic flux. In addition, a differentiator 4b differentiates the flux reference  and multiplies the differential result by L₂/R₂M to obtain a forcing current I FOR . This current I FOR compensates for a delay when the magnetic flux changes. The sum of the forcing current I FOR and the excitation current Io is the flux current component I 1g *. In addition, a slip frequency reference ωs of the induction motor 13 is calculated by dividing the product of the output from the divider 2 and the secondary resistance R₂ by the flux reference . Current detectors 14a, 14b, and 14c detect input currents I u , I v , and I w of the motor 13, respectively. These input currents are decomposed into two-phase DC amounts I 1q-F and I 1d-F by a three-phase/two-phase converter 11. A deviation between the torque current component I 1q * an output from the vector controller 4 and the real current I q-F and that between the flux current component I 1d * and the real current I 1d-F are supplied to a current controller 5 constituted by proportional-plus-integral control elements. The controller 5 independently executes proportional-plus-integral control for the deviation between the torque current component and the real current and that between the flux current component and the real current to obtain voltage references Eq and Ed and outputs the obtained references Eq and Ed to a voltage vector arithmetic operation unit 6. The unit 6 uses the voltage references Eq and Ed to calculate a voltage amplitude reference E and a voltage phase ϑv in accordance with the following arithmetic operations: E = Eq² + Ed² ϑv = tan⁻¹(Eq/Ed) A primary frequency ωo as the sum of the real speed ωr and the slip frequency reference ωs is converted into a primary frequency phase ϑo via an integrator 10, and a voltage phase ϑ from a voltage reference fixed winding is calculated on the basis of the primary frequency phase ϑ and the voltage phase ϑv. The calculated voltage phase ϑ and the above voltage amplitude reference E are supplied to a three-phase reference arithmetic operation unit 7. The unit 7 executes the following arithmetic operations to output the reference voltage signals E u , E v , and E w of three phases U, V, and W, respectively, consisting of sine waves having phases shifted through 120° from each other: E u = Esinϑov E v = Esin(ϑov - 2/3π) E w = Esin(ϑov + 2/3π) The operation of the cycloconverter having the above arrangement will be described below with reference to FIG. 4. The reference voltage signal E u calculated as described above is input to the controller 8. The controller 8 executes asymmetrical control within the range of |ωr| ≦ ωα and symmetrical control within the range of |ωr| ≧ ωβ (> 0). Within the range of ωα ≦ ωr ≦ ωβ, in order to continuously, smoothly switch control for the output voltage in each stage from the asymmetrical to symmetrical control, the controller 8 calculates the reference voltage signals E u 1S and E u 2S on the basis of the flow chart shown in FIG. 2: if E u ≧ 0, E u 1S = (Eα - Vc)sinϑov + Vc E u 2S = E u - E u 1S if E u < 0, E u 1S = E u - E u 2S E u 2S = (Eα - Vc)sinϑov - Vc Since Vc satisfies Vc = Eα within the range of |ωr| ≦ ωα, the reference voltage signals E u 1S and E u 2S have voltage waveforms shown in FIG. 4A. That is, the firing angle of converters 24 and 26 in the first stage are controlled such that their output voltages become the signal E u 1S, and those of converters 23 and 25 in the second stage are controlled such that their output voltages become the signal E u 2S. Therefore, the AC power supplied from the U-phase converter unit to the induction motor 13 has the same sine wave as that of the reference voltage signal E u shown in FIG. 4D. Note that in the asymmetrical control, the firing angles of the converters in the first and second stages of the U-phase converter unit are controlled such that the output voltages from the two stages become the reference voltage signals E u 1S and E u 2S shown in FIG. 4A, respectively. Since Vc satisfies Vc = 0 within the range of |ωr| ≧ ωβ, the reference voltage signals E u 1S and E u 2S have voltage waveforms shown in FIG. 4C. That is, the reference voltage signals E u 1S and E u 2S have the identical symmetrical waveforms, and the firing angles of the converters 24 and 26 in the first stage and the converters 23 and 25 in the second stages are controlled such that their output voltages become E u 1S and E u 2S, respectively. Therefore, the AC power supplied from the U-phase converter unit to the induction motor 13 has the same sine wave as that of the reference voltage signal E u shown in FIG. 4D. Note that the control as described above is the symmetrical control. Vc varies from Vc = Eα to Vc = 0 within the range of ωα ≦ ωr ≦ ωβ. That is, the firing angles of the converters in the first and second stages are controlled so that the output voltages E u 1S and E u 2S in the first and second stages of the U-phase converter unit have voltage waveforms shown in FIG. 4B in accordance with the speed as shown in FIG. 3, thereby obtaining the reference voltage signal E u as a synthetic voltage of the two converter outputs, i.e., a voltage to be supplied to the motor 13. As a result, the asymmetrical control can be switched to the symmetrical control and vice versa without distorting the output voltage to be supplied to the induction motor 13. In addition, a stable operation can be realized in a high-speed operation range. Although the operation has been described above by taking the U-phase output voltage as an example, the output voltage can be similarly determined with respect to the V or W phase. As has been described above, according to the present invention, there is provided a cycloconverter capable of improving the input power factor in a low-speed operation range and realizing a stable operation up to a high-speed operation range without distorting the output voltage and a method of controlling the same. Although the cycloconverter of the present invention is applied to a frequency converting apparatus of an induction motor in the above embodiment, the present invention is not limited to the above embodiment.

Claims: A cycloconverter comprising: a converter group (23 to 26), having at least two cascade-connected converters (23, 24 or 25, 26) for controlling a firing angle of each of said converter (23 to 26) to convert an AC power given to each of said converters into an AC power having an arbitrary frequency; reference voltage generating means (1 to 7) for generating a reference voltage signal having a frequency corresponding to the frequency of the AC power to be output from said converter group (23 to 26); first output voltage generating means (52) for generating a first voltage signal E u 1S obtained by adding a DC voltage Vc to an AC signal (Ea-Vc) sin⊙ov having an amplitude corresponding to a difference between a maximum output voltage Ea of said converter (23 to 26) and the DC voltage VC, i.e. E u 1S = (Ea - Vc) sinΘov + Vc, and a second voltage signal E u 2S obtained by subtracting the first voltage signal E u 1S from the reference voltage signal E u ; i.e. E u 2S = E u - E u 1S second output voltage generating means (53) for generating a third voltage signal E u 2S obtained by subtracting the DC voltage Vc from the (Ea - Vc) sinΘov having an amplitude corresponding to a difference between the DC voltage Vc and the maximum output voltage Ea of said converter (23 to 26), i.e. (Ea - Vc) sinΘov - Vc and a fourth voltage signal obtained by subtracting the third voltage signal E u 2S from the reference voltage signal: E u , i.e. E u 1S = E u - E u 2S, DC component varying means (8) for varying the DC voltage in accordance with the frequency of the reference voltage signal; first firing angle control means (55) for supplying, while the reference voltage signal is positive, a firing signal corresponding to a voltage value of the first voltage signal to one of said converters (24, 26) and a firing signal corresponding to a voltage value of the second voltage signal to the other of said converters (23, 25); and second firing angle control means (55) for supplying, while the reference voltage signal is negative, a firing signal corresponding to a voltage value of the third voltage signal to said the other said converters (23, 25) and a firing signal corresponding to a voltage value of the fourth voltage signal to said one of said converters (24, 26). A cycloconverter according to claim 1, characterized in that said DC component varying means (8) includes means (S2, S3) for continuously decreasing the DC voltage when the frequency of the reference voltage signal changes from a first frequency to a second frequency higher than the first frequency and continuously increasing the DC voltage when the frequency changes from the second frequency to the first frequency. A cycloconverter according to claim 2, characterized in that said DC component varying means (8) includes means (S2, S3) for fixing the DC voltage at the maximum output voltage of said converter when the frequency of the reference voltage signal is lower than the first frequency. A cycloconverter according to claim 2, characterized in that said DC component varying means (8) includes means (S2, S3) for fixing the DC voltage at 0 while the frequency of the reference voltage signal is higher than the second frequency. A cycloconverter according to claim 1, characterized in that said converter group (23 to 26) outputs the frequency-converted AC power to a motor (13), and said DC component varying means (8) varies the DC voltage in accordance with an operation frequency of said motor (13). A cycloconverter according to claim 1, characterized in that said converter group includes a plurality of positive converters (23 to 26) through which a positive component of an AC current supplied to said converter group (23 to 26) flows and plurality of a negative converters (23, 24) through which a negative component of the AC current flows. A cycloconverter according to claim 6, characterized in that each of said positive and negative converters (23 to 26) is constituted by at least one thyristor. A cycloconverter according to claim 1, characterized in that a cyclic current flows from said positive converters (23, 24) to said negative converters (25, 26) in said converter group (23 to 26). A cycloconverter according to claim 1, characterized in that said voltage reference generating means (1 to 7) includes flux arithmetic operating means (3) for calculating a flux reference of a motor (13) which receives the AC power output from said converter group (23 to 26) in accordance with a real speed of the motor (13), means (1) for calculating a torque reference of the motor (13) in accordance with an externally supplied speed reference and the real speed, means (4) for calculating a torque current component and a flux current component in accordance with the flux reference and the torque reference, means (11) for detecting a real current as an input current to the motor (13) and converting the detected current into two-phase real currents, means (5) for executing proportional-plus-integral control for a deviation between one of the real currents generated by said converting means (11) and the torque current component and a deviation between the other real current generated by said converting means (11) and the flux current component and outputting an arithmetic operation result signal, means (6) for calculating a voltage amplitude reference and a voltage phase in accordance with the arithmetic operation result signal output from said proportional-plus-integral control means (6), means for calculating a primary frequency phase in accordance with the real speed and a slip frequency reference of said motor (13), means for calculating a phase of the voltage reference in accordance with the voltage phase and the primary frequency phase, and means (11) for calculating the voltage reference in accordance with the phase calculated by said phase calculating means and the voltage amplitude reference. A method of controlling a cycloconverter, for supplying an AC power having a given frequency to a converter group (23 to 26) having at least two cascade-connected converters (23, 24 or 25, 26) and controlling a firing angle of each of said converters (23 to 26), thereby converting the AC power supplied to said converter group (23 to 26) into an AC power having a predetermined frequency, characterized by comprising the steps of: controlling an output voltage from one of said converters (24, 26) of said converter group (23 to 26) in accordance with the polarity of the reference voltage signal having a predetermined frequency to obtain an AC output voltage E u 1S including a synthetic voltage including an AC voltage (Ea - Vc) sinΘov and a predetermined DC voltage VC, said AC voltage having an amplitude equal to a difference between a maximum output voltage Ea of said converter and the DC voltage Vc, i.e. E u 1S = (Ea - Vc) sinΘov + Vc, controlling an output voltage E u 2S from the other of said converters (23, 25) to obtain an AC output voltage corresponding to a difference between the output voltage E u 1S from said one of said converters (24, 26) and the reference voltage signal (Eu); and continuously varying a voltage value of the DC voltage in accordance with the frequency of the reference voltage signal. A method according to claim 10, characterized by further comprising: a asymmetrical control step of controlling the output voltage from said one of said converters (24, 26) of said converter group (23 to 26) to be the maximum output voltage of said one of said converters (24, 26), and controlling the output voltage from said the other of said converters (23, 25) of said converter group (23 to 26) to be the AC output voltage obtained by subtracting the maximum output voltage from the reference voltage signal; a symmetrical control step of controlling the output voltage from each of said converters (23 to 26) to one and be a voltage value about 1/2 that of the reference voltage signal; and a switching step of executing switching between said asymmetrical and symmetrical control steps in accordance with a distortion in the output voltage supplied from said converter group (23, 26). A method according to claim 10, characterized by further comprising steps of: generating a reference signal having a frequency corresponding to a frequency of the AC power to be output from said converter group (23 to 26); generating a first voltage signal obtained by adding a DC voltage to an AC signal having a difference between the maximum output voltage of said converter (23 to 26) and the DC voltage as an amplitude thereof and a second voltage signal obtained by subtracting the first voltage signal from the reference voltage signal; generating a third voltage signal obtained by subtracting the DC voltage from the AC signal having a difference between the DC voltage and the maximum output voltage of said converter (23 to 26) as an amplitude thereof and a fourth voltage signal obtained by subtracting the third voltage signal from the reference signal; varying the DC voltage in accordance with the frequency of the reference voltage signal; supplying, while the reference voltage signal is positive, a firing signal corresponding to a voltage value of the first voltage signal to said one of said converters (24, 26) and a firing signal corresponding to a voltage value of the second voltage signal to said the other of said converters (23, 25); and supplying, while the reference voltage signal is negative, a firing signal corresponding to a voltage value of the third voltage signal to said the other of converters (23, 25) and a firing signal corresponding to a voltage value of the fourth voltage to said one of said converters (24, 26).

Inventors: OKAYASU MASAMI C O INTELLECTUA, TATARA SHINJI C O INTELLECTUAL, OKAYASU, MASAMI, C/O INTELLECTUAL PROPERTY DIV., TATARA, SHINJI, C/O INTELLECTUAL PROPERTY DIV.
Applicants: TOSHIBA KK, KABUSHIKI KAISHA TOSHIBA
Application year: 1991
Main classification: H02M5
IPC codes: H02P27, H02M5