Costas loop carrier wave reproducing circuit

Abstract: A Costas loop carrier wave reproducing circuit includes a first differential amplifier circuit, a second differential amplifier circuit, a multiplying circuit, a first square circuit, a second circuit, and a double-balanced differential amplifier circuit. The first differential amplifier circuit obtains an inphase output from an inphase demodulated signal obtained by performing synchronous detection of a component in phase with a four-phase modulated wave. The second differential amplifier circuit obtains an orthogonal output from an orthogonal demodulated signal obtained by performing synchronous detection of an orthogonal component of the four-phase modulated wave. The multiplying circuit multiplies outputs from the first and second differential amplifier circuits. The first square circuit obtains a square output of an output from the first differential amplifier circuit. The second square circuit obtains a square output of an output from the second differential amplifier circuit. The double-balanced differential amplifier circuit obtains an output difference between the first and second square circuits and outputting a product of the output difference and an output from the multiplying circuit as a PLL control signal.

Description: Background of the Invention The present invention relates to a demodulator for a four-phase modulated wave and, more particularly, to a Costas loop carrier wave reproducing circuit. In general, a digital phase modulation scheme for carrier waves, called "PSK", is superior in its frequency band characteristics and code error rate characteristics to other modulation schemes such as amplitude modulation, frequency modulation, and pulse modulation, and hence is widely used in, e.g., PCM microwave communications, satellite communications, and data transmission modems. This phase modulation scheme includes two-, four-, and eight-phase PSK schemes which can transmit 1-, 2-, and 3-bit data per one sampling period, respectively. The four-phase PSK scheme is most widely used in practice because a PSK scheme for more than four phases demands strict identification characteristics and the arrangement of the modulator becomes complicated. In this four-phase PSK scheme, the digital code of a signal to be transmitted is divided for every two bits, and a carrier wave is modulated by a phase shift amount corresponding to one of the four combinations of 2-bit data. In many four-phase schemes currently used in practice, phase shift amounts of 0, π/2, π, and 3π/2 respectively correspond to the four combinations (0, 0), (0, 1), (1, 0), and (1, 1). According to this scheme, a 2-bit signal component is contained in an inphase component Pcoswct and an orthogonal component Qsinwct of a carrier wave. For this reason, in demodulation, the inphase and orthogonal components of the carrier wave are respectively detected to identify their polarities. In this case, a synchronous detection scheme is often used as a detection scheme because it has good code error rate characteristics. Unlike other delay detection schemes, however, in the synchronous detection scheme, a carrier wave reproducing circuit must be prepared on the reception side to generate a carrier wave having a correct phase. This carrier wave reproducing circuit is designed to control a phase-locked loop circuit (PLL) by using a signal obtained by removing a code component from an input carrier wave. Fig. 3 is a block diagram showing the principle of a four-phase demodulating circuit. Referring to Fig. 3, reference numerals 1, 2, and 7 denote multipliers; 3, 4, and 8, LPFs (low-pass filters); 5, an adder; 6, a subtracter; 9, a VCXO; 10, a phase shifter; and 20, Costas loop carrier wave reproducing circuit. A signal S as a four-phase modulated wave is represented by the following equation, provided that wt is the frequency of a carrier wave, and  is a phase shift of 0, 90°, 180°, or 270°: S = Ecos(wt + ) If two outputs (one of them is delayed by π/2) from the VCXO (voltage-controlled crystal oscillator) 9 are respectively represented by A and B, the following equations can be established: A = E 1 sin( wt +  1 ) B = E 1 cos( wt +  1 ) When the four-phase modulated wave S and the oscillator outputs A and B are demodulated by the multipliers 1 and 2 and are filtered by the LPFs (low-pass filters) 3 and 4, the multiplication of equations (1) and (2) is rewritten as follows: Ecos(wt + ) x E 1 sin( wt +  1 ) = E · E 1 2 {sin(2 wt +  +  1 ) - sin( 1 - )} If this equation is filtered by an LPF, the following expression is obtained: ≒ E · E 1 2 sin( -  1 ) Similarly, the multiplication of equations (1) and (3) is represented by E cos( wt + ) x E 1 cos( wt +  1 ) = E · E 1 2 {cos(2 wt +  +  1 ) + cos( -  1 )} ≒ E · E 1 2 cos( -  1 ) The outputs from the LPFs are then processed by the Costas loop carrier wave reproducing circuit 20. The circuit 20 serves to keep the phase of an output from the VCXO constant regardless of the phase state (0. 90°, 180°, 270°) of an input four-phase modulated wave. If the demodulated signals of equations (4) and (5) are respectively represented by P and Q, P + Q is output from an output E point of the adder 5; P - Q, from an output F point of the subtracter 6; and P x Q x (P + Q) x (P - Q), from an output G point of the multiplier 7. At the output E, the following equations are established: P x Q = E · E 1 2 sin( -  1 ) x E · E 1 2 cos( -  1 ) = ( E · E 1 ) 2 8 sin(2 - 2 1 ) ( P + Q ) X ( P - Q ) = P 2 - Q 2 = { E · E 1 2 sin( -  1 )} 2 - { E · E 1 2 cos( -  1 )} 2 = - ( E · E 1 ) 2 4 cos(2 - 2 1 ) At the output G point, the following equation can be obtained by the multiplication of equations (6) and (7): ( P + Q ) · ( P - Q ) · PQ = ( E · E 1 ) 2 8 sin (2 - 2 1 ) x { - ( E · E 1 ) 2 4 cos(2 - 2 1 )} = - ( E · E 1 ) 4 64 {sin(4 - 4 1 ) + sin} = - ( E · E 1 ) 4 64 sin4 ( -  1 ) In this case, a phase difference ( - ₁) between the four-phase modulated wave, the output signal from the VCXO is nπ/2, and the output voltage relatively becomes 0 (n = integer), and the PLL is locked in this state. If, therefore, this output is fed back to the VCXO 9 through the LPF 8, the data of the signals P and Q are demodulated. Fig. 2 is a circuit diagram showing the detailed arrangement of the conventional Costas loop carrier wave reproducing circuit 20 shown in Fig. 3. This conventional technique is disclosed in U.S.P. No. 4,694,204, Sep. 15, 1987, assigned to NEC Corporation. This circuit comprises a first multiplying circuit constituted by a double differential amplifier circuit consisting of transistors Q₂₃ to Q₂₈ and a constant current source I₈, a second multiplying circuit constituted by a double differential amplifier circuit consisting of transistors Q₂₉, to Q₃₂ for providing collector currents to the first multiplying circuit as common emitter currents, a third multiplying circuit constituted by a double differential amplifier circuit consisting of transistors Q₃₃ to Q₃₆ for providing collector currents to the second multiplying circuit as common emitter currents, load resistors R₁₁ and R₁₂ for applying a collector current, as an output voltage, from the fourth multiplying circuit to output terminals a and b , three differential amplifiers, constituted by transistors Q₁, Q₂, Q₂₁, Q₂₂, Q₃₇, and Q₃₈, constant current sources I₁, I₇, and I₉, and load resistors R₁₃ to R₁₆, for generating input voltages of the demodulated signals P and Q and their sum and difference signals, a bias circuit, constituted by DC power sources V₁ and V₂ and resistors R₉ and R₁₀, for biasing the three differential amplifiers, and a peripheral circuit constituted by DC power sources V₃, V₄, and V₅. Assume that the inphase demodulated signal P is input through a coupling capacitor C₁. In this case, since the bases of the transistors Q₁ and Q₃₈ are biased by the bias power source V₁ through the bias resistor R₉, and the bases of the transistors Q₁ and Q₃₈ are directly biased by the bias power source V₁, a signal in phase with the inphase demodulated signal P appears at each of the load resistors R₁₃ and R₁₄. Similarly, if the orthogonal demodulated signal Q is input through a coupling capacitor C₂, signals in phase with and in opposite phase to the demodulated signal Q appear at the load resistors R₁₅ and R₁₆ of the differential amplifier constituted by the transistors Q₂₁ and Q₂₂. In the first multiplying circuit, the inphase demodulated signal P is input to the base of the transistor Q₂₃, and the base of the transistor Q₂₄ is biased by the DC power source V₁, while a constant current I₀ from the constant current source I₈ flows in the commonly connected emitter. In this state, a current I C0 and a current I C0 which are in phase with and in opposite phase to the demodulated signal P respectively flow in the collectors of the transistors Q₂₃ and Q₂₄. In addition, the bases of the transistors Q₂₅ and Q₂₈ of the double amplifier circuit are biased by the DC power source V₂, and the orthogonal demodulated signal Q is input to the bases of the transistors Q₂₆ and Q₂₇, while the commonly connected emitter is controlled by the currents I C0 and I C0 which are in phase with and in opposite phase to the demodulated signal P. Therefore, the sum of the collector currents of the transistors Q₂₅ and Q₂₇ and the sum of the collector currents of the transistors Q₂₆ and Q₂₈ become currents I C1 and I C2 corresponding to the product output (P x Q) of the two demodulated signals and an output in opposite phase thereto. Subsequently, the product output currents I C1 and I C2 of the demodulated signals P and Q respectively become common emitter currents flowing in the double amplifier circuit constituting the second multiplying circuit. In the double differential amplifier circuit of the second multiplying circuit, signals in phase with the demodulated signals P and Q are respectively supplied to the bases of the transistors Q₂₉ and Q₃₂ and to the bases of the transistors Q₃₀ and Q₃₁ in a differential manner through the load resistors R₁₃ and R₁₅. In this state, collector output currents I C3 and I C4 from the second multiplying circuit are proportional to values obtained by multiplying the product output (P x Q) of the two demodulated signals P and Q by the signal (P - Q). In the same manner as described above, these two collector output currents I C3 and I C4 become common emitter currents flowing in the double differential amplifier circuit constituting the third multiplying circuit. Since a signal in phase with the demodulated signal P and a signal in opposite phase to the demodulated signal Q are differentially supplied to the bases of the transistors Q₃₃ to Q₃₆ of the double differential amplifier circuit of the third multiplying circuit through the load resistors R₁₄ and R₁₆, collector output currents I C5 and I C6 from the third multiplying circuit correspond to values obtained by multiplying the outputs from the second multiplying circuit by the signal {P - (-Q)}, i.e., the sum signal (P + Q). With this operation, a voltage proportional to a voltage obtained by multiplying all the four signals, i.e., the demodulated signals P and Q, the sum signal (P + Q), and the difference signal (P - Q), as the multiplication result obtained by the first to third multiplying circuits, is output across output terminals a and b . In this conventional Costas loop carrier wave reproducing circuit, the three double differential amplifier circuits are vertically stacked on each other to calculate P x Q x (P + Q) x (P - Q). With this arrangement, in order to linearly operate the circuit, 4 x (VBE + amplitude) + a voltage drop due to the load resistors + a voltage required to normally operate the constant current sources is required as a power source voltage. In practice, however, if VBE = 0.75 V, an amplitude = 0.5 V PP , a voltage drop due to the load resistors = 1 V, and a voltage applied to the constant current sources = 1 V, V CC > (0.75 + 0.5) x 4 + 1 +1 = 7 V is required. Therefore, the circuit cannot be operated by a power source voltage of 5 V. Summary of the Invention It is an object of the present invention to provide a Costas loop carrier wave reproducing circuit which allows a decrease in power source voltage. It is another object of the present invention to provide a Costas loop carrier wave reproducing circuit which allows the use of a MOS transistor. In order to achieve the above objects, according to the present invention, there is provided a Costas loop carrier wave reproducing circuit comprising a first differential amplifier circuit for obtaining an inphase output from an inphase demodulated signal obtained by performing synchronous detection of a component in phase with a four-phase modulated wave, a second differential amplifier circuit for obtaining an orthogonal output from an orthogonal demodulated signal obtained by performing synchronous detection of an orthogonal component of the four-phase modulated wave, a multiplying circuit for multiplying outputs from the first and second differential amplifier circuits, a first square circuit for obtaining a square output of an output from the first differential amplifier circuit, a second square circuit for obtaining a square output of an output from the second differential amplifier circuit, and a double-balanced differential amplifier circuit for obtaining an output difference between the first and second square circuits and outputting a product of the output difference and an output from the multiplying circuit as a PLL control signal. Brief Description of the Drawings Fig. 1 is a circuit diagram showing a detailed arrangement of a Costas loop carrier wave reproducing circuit according to an embodiment of the present invention; Fig. 2 is a circuit diagram showing a detailed arrangement of a conventional Costas loop carrier wave reproducing circuit; and Fig. 3 is a block diagram showing the principle of a four-phase demodulating circuit. Description of the Preferred Embodiment The present invention will be described below with reference to the accompanying drawings. Fig. 1 shows a detailed arrangement of a Costas loop carrier wave reproducing circuit according to an embodiment of the present invention. This circuit comprises: a first differential amplifier circuit constituted by transistors Q₁ and Q₂, a constant current source I₁, and load resistors R₁ and R₂; a multiplying circuit constituted by transistors Q₃ to Q₁₀, constant current sources I₂ and I₃, and load resistors R₃ and R₄; a first square circuit constituted by transistors Q₁₁ and Q₁₂, and a constant current source I₄; a second square circuit constituted by transistors Q₁₉ and Q₂₀ and a constant current source I₆; a second differential amplifier circuit constituted by transistors Q₂₁ and Q₂₂, a constant current source I₇, and load resistors R₇ and R₈; and a double-balanced differential amplifier circuit constituted by transistors Q₁₃ to Q₁₈, a constant current source I₅, and load resistors R₅ and R₆. Assume that an inphase demodulated signal P is input through a coupling capacitor C₁. Since the base of the transistor Q₁ is biased by a bias power source V₁ through a bias resistor R₉ while the base of the transistor Q₂ is directly biased by the bias power source V₁, signals in phase with and in opposite phase to the demodulated signal P respectively appear at the load resistors R₁ and R₂. Similarly, if an orthogonal demodulated signal Q is input through a coupling capacitor C₂, signals in phase with and in opposite phase to the demodulated signal Q respectively appear at the load resistors R₇ and R₈ of the differential amplifier circuit constituted by the transistors Q₂₁ and Q₂₂. In the multiplying circuit constituted by the transistors Q₃ to Q₁₀, a signal which is produced across the load resistor R₂ and is in phase with the demodulated signal P is input to the common base of the transistors Q₃ and Q₄, a signal which is produced across the load resistor R₁ and is in opposite phase to the demodulated signal P is input to the common base of the transistors Q₉ and Q₁₀, a signal which is produced across the load resistor R₇ and is in phase with the demodulated signal Q is input to the common base of the transistors Q₅ and Q₈, and a signal which is produced across the load resistor R₈ and is in opposite phase to the demodulated signal Q is input to the common base of the transistors Q₆ and Q₇. In this state, the product of the demodulated signals P and Q appear at the load resistors R₃ and R₄. In the first square circuit constituted by the transistors Q₁₁ and Q₁₂, a signal which is produced across the load resistor R₂ connected to a power source V₂ lower than a power source V₄ and is in phase with the demodulated signal P is level-shifted and input to the base of the transistor Q₁₁, and a signal which is produced across the load resistor R₁ connected to the power source V₂ and is in opposite phase with the demodulated signal P is level-shifted and input to the base of the transistor Q₁₂. In this state, the square of the demodulated signal P is obtained as an output from the common emitter of the transistors Q₁₁ and Q₁₂. Similarly, in the square circuit constituted by the transistors Q₁₉ and Q₂₀, a signal which is produced across the load resistor R₈ connected to the power source V₂ and is in opposite phase to the demodulated signal Q is level-shifted and input to the base of the transistor Q₁₉, and a signal which is produced across the load resistor R₇ connected to the power source V₂ and is in phase with the demodulated signal Q is level-shifted and input to the base of the transistor Q₂₀. In this state, the square of the demodulated signal Q is obtained as an output from the common emitter of the transistors Q₁₉ and Q₂₀. In the double-balanced differential amplifier circuit constituted by the transistors Q₁₃ to Q₁₈, an output signal corresponding to the square of the demodulated signal P is input to the base of the transistor Q₁₃, and an output signal corresponding to the square of the demodulated signal Q is input to the base of the transistor Q₁₄. As a result, a current corresponding to the difference between a demodulated signal P² and a demodulated signal Q² flows in the collectors of the transistors Q₁₃ and Q₁₄. Meanwhile, the output signal obtained as the square of the demodulated signals P and Q is level-shifted and input to the bases of the transistors Q₁₅ and Q₁₈ constituting the double-balanced differential amplifier circuit through a load resistor R₃ connected to a power source V₃ lower than the power source V₄. Similarly, the output signal is level-shifted and input to the bases of the transistors Q₁₆ and Q₁₇ through a load resistor R₄ connected to the power source V₃. As a result, in the double-balanced differential amplifier circuit, the result obtained by performing a multiplication of the demodulated signals P and Q according to P x Q x (P² - Q²) is output to the load resistors R₅ and R₆. The square circuit constituted by the transistors Q₁₁, Q₁₂, Q₁₉, and Q₂₀ will be described in detail below. If the common emitter voltage of the transistors Q₁₁ and Q₁₂ is represented by V E ; the collector currents of the transistors Q₁₁ and Q₁₂, I C1 and I C2 ; the thermal voltage of the transistors Q₁₁ and Q₁₂, v T ; a reverse saturation current, I S ; and the base bias voltage of the transistors Q₁₁ and Q₁₂, V A , the following equations can be obtained: V A + P - V E = v T ℓ n I C1 I S V A - P - V E = v T ℓ n I C2 I S If equations (1) and (2) are added together, the following equation is established: 2 V A - 2 V E = v T ℓ n I C1 I C2 I S 2 If the mutual conductance of the differential circuit is represented by gm, and the current supplied from the constant current source I₄ is represented by 2I₀ (I₀ >> P x gm), the following formulae are obtained: I C1 ≒ I 0 + 2 x P x gm I C2 ≒ I 0 - 2 x P x gm Therefore, equation (3) is rewritten as follows: 2 V A - 2 V E = v T ℓ n { I O +2P x gm I O -2P x gm I S 2 = v T ℓ n { I O 2 I S 2 (1 - 4 P 2 gm 2 I O 2 )} Since I O >> P x gm, 2 V A - 2 V E ≒ 2 v T ℓ n I O I S - v T 4 x P 2 x gm 2 I O 2 If gm is expressed by v T and I O , gm = 2 I O 4 v T = I O 2 v T Therefore, V E is given by V E = P 2 2 v T + V A - v T ℓ n I O I S With this operation, an output proportional to the square of the input signal P can be obtained. As has been described above, according to the present invention, a practical power source voltage can be noticeably decreased as compared with the conventional circuit. Assume that the base-emitter voltage of a transistor is 0.75 V, the amplitude is 0.5 V PP , the voltage drop due to the load resistance of an output is 1 V, and the voltage applied to a constant current source is 1 V. In this case, the power source voltage is V BE . + 0.5 V x 2 + 1 V + 1 V = 3.75 V. Therefore, even if the power source voltage is set to be 5 V, the circuit can be easily operated. That is, even a MOS transistor can be used.

Claims: A Costas loop carrier wave reproducing circuit comprising: a first differential amplifier circuit (Q₁, Q₂, I₁, R₁, R₂) for obtaining an inphase output from an inphase demodulated signal obtained by performing synchronous detection of a component in phase with a four-phase modulated wave; a second differential amplifier circuit (Q₂₁, Q₂₂, I₇, R₇, R₈) for obtaining an orthogonal output from an orthogonal demodulated signal obtained by performing synchronous detection of an orthogonal component of the four-phase modulated wave; a multiplying circuit (Q₃, Q₄, Q₅, Q₆, Q₇, Q₈, Q₉, Q₁₀, I₂, I₃, R₃, R₄) for multiplying outputs from said first and second differential amplifier circuits, characterised by: a first square circuit (Q₁₁, Q₁₂, I₄) for obtaining a square output of an output from said first differential amplifier circuit; a second square circuit (Q₁₉, Q₂₀, I₆) for containing a square output of an output from said second differential amplifier circuit; and a double-balanced differential amplifier (Q₁₃, Q₁₄, Q₁₅, Q₁₆, Q₁₇, Q₁₈, I₅, R₅, R₆) circuit for obtaining an output difference between said first and second square circuits and outputting a product of the output difference and an output from said multiplying circuit as a PLL control signal. A circuit according to claim 1, further comprising a low-pass filter (8) for extracting a low-frequency component from an output of said double-balanced differential amplifier circuit, and a voltage-controlled oscillator (9) whose output frequency is controlled by an output voltage from said low-pass filter to generate inphase (A) and orthogonal (B) reference phase carrier waves for synchronous detection. A circuit according to claim 1, wherein said multiplying circuit comprises a first transistor (Q₃, Q₄) for receiving a first input as a base input, a second transistor (Q₉, Q₁₀) for receiving a second input in opposite phase to the first input as a base input, a third transistor for receiving a third input as a base input, a fourth transistor (Q₇, Q₈) connected to said third transistor to constitute a double differential circuit, for receiving a fourth input in opposite phase to the third input as a base input, a first constant current (I₂) source to which emitters of said first and third transistors are commonly connected, a second constant current source (I₃) to which emitters of said second and fourth transistors are commonly connected, and load resistors (R₃, R₄) to which collectors of said third and fourth transistors constituting said double differential circuit are commonly connected. A circuit according to claim 3, wherein said first and second transistors are respectively constituted by transistor pairs having collectors, bases, and emitters, all of which are commonly connected, and said third and fourth transistors are respectively constituted by transistor pairs having bases and emitters, all of which are commonly connected, and collectors respectively connected to two load resistors. A circuit according to claim 1, wherein a load resistor (R₃) of said multiplying circuit is connected to a power source having a voltage (V₃) lower than a further power source voltage (V₄) to supply a level-shifted output to said double-balanced amplifier circuit. A circuit according to claim 1, wherein said first and second differential amplifier circuits have load resistors (R₁, R₂; R₇, R₈) connected to a power source (V₂) having a voltage lower than a further power source voltage (V₄), and respectively supply level-shifted outputs to said multiplying circuit and said square circuits.

Inventors: NAKAYAMA ISAO, NAKAYAMA, ISAO
Applicants: NIPPON ELECTRIC CO, NEC CORPORATION
Application year: 1991
Main classification: H04L27
IPC codes: H04L27