Advance/retard control circuit with PDM accumulator and second order loop filter

Abstract: A digital loop filter translates a multi-bit phase error input into a high resolution control signal utilizable as an advance-retard control for a multi-phase clock generator. The digital filter couples the multi-bit phase error input to the clock generator via a pulse density modulation (PDM) accumulator, providing multi-phase adjustment in a single sample clock cycle based on the overflow or underflow of the PDM accumulator. Variable PDM cycles are used to control loop filter bandwidth, permitting adjustable capture sequences. Thus, real proportional control of the multi-phase clock generator is limited only by the word size of the phase error input.

Description: The present invention relates to phase locked loop systems and, in particular, to a pulse density modulation technique for translating a multi-bit phase error input into a high resolution control signal for a stepped multi-phase clock generator. A phase locked loop (PLL) is a frequency-selective feedback system which synchronizes with an input signal and then tracks changes in input signal frequency. As shown in the Fig. 1, a basic analog PLL 10 includes three primary elements: a phase detector, a loop filter with some gain, and a voltage controlled oscillator (VCO). The frequency of the input signal to the PLL 10 has an instantaneous phase  i (s). The instantaneous phase of the VCO output frequency is  o (s). The input and output phases are compared by the phase detector to provide a phase error  e (s). The phase error  e (s) can be expressed in Laplace notation as  e (s) =  i (s) M -  o (s) N V d (S) = K d  e (S) where K d (volts per radian) is the gain constant of the phase detector. The output voltage of the phase detector is filtered by the loop filter utilizing a loop transfer function F(s) which rejects out-of-band noise and high-frequency signal components. The output voltage of the loop filter is given by V 2 (s) = V d (s)F(s) = K d F(s) e (s) The corresponding change in the output frequency of the VCO is ΔW = K 2 V 2 (s) where K 2 (radians per second per volt) is the gain constant of the VCO. Since ΔW = d o (s)/dt = s 2 (s)  o (s) = K o F(s) s Combining the above equations provides the basic loop transfer function  o (s)  i (s) = H(s) = K o F(s) s+K o F(s) = KF(s) s+KF(s) where K = K 2 K d . Also, as stated above  e (s) =  i (s) -  o (s) Therefore,  e (s)  i (s) = I· s s+KF(s) The PLL 10 behaves like any feedback system. For proper operation of the loop 10, three parameters must be chosen independently, depending upon the application: (1) the natural frequency W n of the loop, (2) a damping factor ζ, and (3) the DC loop gain K v = KF(o), where F(o) is the DC gain of the loop filter 14. For a basic first-order PLL, the loop filter is omitted. Thus, F(S) = 1 and the basic loop transfer function becomes H(s) = K s + k Since the only variable in a first-order loop is K v = K, the usefulness of a first-order loop is very limited. In many applications, it is desirable to utilize a second-order loop filter configured as shown in Fig. 2, which illustrates an active filter using an operational amplifier. For this configuration, the loop transfer function becomes H(s) = 2ζW n s s 2 + 2ζW n s + W n 2 where W 2 n = K T i ζ = T 2 2 W n (T 1 = CR 1 ; T 2 = CR 2 ) The active filter transfer function is given by F(s) = ST 2 + 1 ST 1 = T 2 T 1 + 1 ST 1 = R 2 R 1 + 1 SCR 1 This is referred to as a proportional-plus-integral control, since the transfer function F(s) comprises the sum of a term which is proportional to the phase error and a term which represents the integration or accumulation of the phase error over time. As shown in Fig. 3, the transfer function F(s) may be realized by two active filters operating in parallel, one for the proportional term R 2 /R 1 and one for the integral term 1/SCR 1 . The proportional and integral terms are then added to provide the control signal for the VCO 16. Both of these latter configurations permit independent choice of W n , ζ, and K v , and are widely used in practical PLL designs. The active filter has the added advantage that the presence of the amplifier makes the DC loop gain K v very high compared to that obtainable with the passive configuration. The active second-order loop filter is, therefore, the most attractive choice for most applications. A digital PLL (DPLL) is a discrete time version of the above-described analog PLL. In a DPLL, the phase error is sampled and quantized in an analog-to-digital (A/D) converter and then processed in a digital discrete time filter. The discrete output of the filter is converted into analog samples by a digital/analog converter and then held in a zero-order hold circuit the output of which controls the VCO. A simpler type of digital PLL, sometimes referred to as a digital phase synchronizer, can be used to provide a stepping signal for advancing or retarding a multi-phase clock generator or digitally controlled clock. In this type of PLL, the sign of the discrete phase error value generated by the A/D converter determines whether the control output of the PLL will advance or retard the stepped clock generator. That is, if during a cycle of the sample clock generated by the clock generator, the input phase advances relative to the phase of the sample clock, then an advance signal is provided to the clock generator to cause a forward phase jump. Conversely, if the input phase lags relative to the phase of the sample clock output of the clock generator, then a retard signal is provided to cause a backward phase jump. A major problem associated with this so-called "bang-bang" approach to phase clock control is that the single-bit sign value utilized to drive the advance/retard signal can only implement single-phase movement of the clock generator in a given cycle of the sample clock and, therefore, does not provide the fine resolution required in high speed data recovery applications. The present invention as defined in the patent claim is suited to overcome this drawback, the feature cited in its preamble being known from GB-A-22 01 859. Preferably, the present invention provides a pulse density modulation (PDM) accumulator, second-order loop filter system that translates a multi-bit phase error input to a high resolution control signal utilizable as an advance/retard control for a multi-phase clock generator (or stepped digitally controlled clock). That is, a digital loop filter couples the multi-bit phase error input to the stepped clock generator via a PDM accumulator, thus providing for multi-phase adjustment of the clock generator within a single cycle of the sample clock output. Variable PDM cycles are used to control loop bandwidth, permitting a high speed initial capture sequence which then can be slowed down as the loop locks on the incoming signal. Use of a digital integrator allows the selection of a large number of discrete values of frequency that will not change, as in conventional RC integrators, due to charge leakage. This allows tracking and retransmission of incoming data using a low cost reference crystal. Real proportional control of the clock generator is limited only by the word size of the phase error input. A PDM accumulator second-order loop filter system in accordance with the present invention comprises a proportional accumulator responsive to the multi-bit phase error input signal for generating a multi-bit proportional term, an integrator accumulator responsive to the proportional term for iteratively accumulating the proportional term to provide an integral term, integrator-plus-proportional means for summing the proportional term and the integral term to provide a multi-bit integrator-plus-proportional term, and a PDM accumulator for accumulating the multi-bit integrator-plus-proportional term to provide a PDM accumulator control signal indicative of both the number of phase jumps to be taken by a stepped clock generator responsive to the PDM control signal and the direction of the phase jumps. A better understanding of the features and advantages of the present invention will be obtained by reference to the following detailed description and accompanying drawings which set forth an illustrative embodiment in which the principles of the invention are utilized. Figure 1 is a block diagram illustrating a basic analog phase locked loop. Figure 2 is a schematic diagram illustrating an active filter of the type utilizable in a conventional analog phase locked loop. Figure 3 is a schematic diagram illustrating the realization of the active filter shown in Fig. 2 utilizing two active filters in parallel to generate proportional and integral terms. Figure 4 is a block diagram illustrating a receiver system that utilizes a PDM accumulator, second-order loop filter system in accordance with the present invention to recover digital data from an analog input signal. Figure 5 is a block diagram illustrating an embodiment of a PDM accumulator, second-order loop filter system in accordance with the present invention. Figure 6 is a map for Figures 6A-6I. Figures 6A-6I combine to provide a schematic diagram illustrating a circuit embodiment of a PDM accumulator, second-order loop filter system in accordance with the present invention. Figure 7 is a diagram illustrating time slot allocation for the PDM accumulator of the system shown in Figs. 6A-6I for different bandwidth settings. Fig. 4 shows a simple block diagram of a receiver system for recovering digital data from an analog input signal. As shown in Fig. 4, an incoming analog modulated carrier waveform received from a communications channel is first processed by a analog front end 10 in the conventional manner; this initial processing typically includes prefiltering of the received signal and automatic gain control. The output signal 12 of the analog front end 10 is then provided to an analog-to-digital (A/D) converter 14. The A/D converter 14 relies on a sample clock signal 16 to recover a digitized replica 18 of the analog input signal from the output of the analog front end 10. This digitized replica 18 is then processed by a digital signal processor (DSP) 20 in accordance with a dedicated DSP algorithm to recover the incoming digital data. The DSP 20 also generates a multi-bit phase error signal 22 which is representative of the phase difference between the incoming signal and the current sample clock output 16 of a multi-phase clock generator or digitally controlled clock 24. As stated above in the Background of the Invention section of this patent document, in a conventional digital phase locked loop, the sign of the digital phase error signal 22, i.e. a single bit signal, is utilized to either advance or retard the phase of the sample clock 16 provided by the multi-phase clock generator 24. Adjustment of the multi-phase clock generator 24 in this manner is limited to a single phase jump per cycle of the sample clock 16. However, in accordance with the present invention, the multi-bit phase error signal 22 is provided to a pulse density modulation (PDM) accumulator, second-order digital loop filter system 100 that, as described in greater detail below, translates the multi-bit phase error signal 22 into a high resolution, control signal utilizable by the multi-phase stepped clock generator 24. The control signal provided by the PDM accumulator system 100 includes components indicative of both the number of phase jumps to be taken by the stepped clock generator 24 in a single cycle of the sample clock 16 and the direction of the phase jumps. Thus, real proportional control of the multi-phase clock generator 24 is limited only by the word size of the phase error input signal 22 to the loop filter 100. An example of a stepped clock generator 24 which may be utilized in practicing the present invention is disclosed in copending and commonly assigned U.S. Patent Application Serial No. 369,474 --now US-A-5,018,169-- filed by Hee Wong et al on June 21, 1989 for a High Resolution Sample Clock Generator with Deglitcher; the just-referenced Wong et al application is hereby incorporated by reference to provide additional background information regarding the subject invention. Fig. 5 shows a block diagram of a PDM accumulator, second-order loop filter system 100 in accordance with the present invention. The system 100 shown in Fig. 5 includes a proportional accumulator 102 which receives a 16-bit phase error input signal from the DSP 20 (Fig. 4). The 16-bit phase error input signal is stabilized in an input register 104 and then divided by divider 106 to provide a 16-bit phase error term proportional to the phase error input signal. The 16-bit proportional term is provided both to an integrator accumulator 108 as well as to integrator-plus-proportional summing circuitry 110. The integrator accumulator 108, which includes a 32-bit adder 112 and a 32-bit register 114, continually accumulates the 16-bit proportional term in the conventional manner to provide an integral term to the summing circuitry 110. The integral-plus-proportional summing circuitry 110, which includes a 16-bit adder 116 and 16-bit register 118, receives the proportional term from the proportional filter 102 and the integral term from the integrator accumulator 108 and sums them in the conventional manner to provide a corresponding 16-bit integrator-plus-proportional term. That is, the 16-bit phase error input signal from the proportional filter 102 is provided to the 16-bit adder 116. The 16-bit adder 116 also receives the 16 most significant bits (MSB) of 32-bit output of the integrator accumulator 108. The 16-bit output of the adder 116 is allowed to stabilize in the 16-bit register 118 which then provides the 16-bit integral-plus-proportional output term to a pulse density modulation (PDM) accumulator 120. As described in greater detail below, the PDM accumulator 120 interfaces the 16-bit integral-plus-proportional term to the multi-phase clock generator 24 (Fig. 4). The PDM accumulator 120 accumulates the integral-plus-proportional term, using a 16-bit adder 122, either 3, 6, 12 or 24 times per period of the sample clock 16 (Fig. 4), depending on a bandwidth control setting. Each time an overflow or underflow of adder 122 occurs, the multi-phase clock generator 24 is enabled via the shift/idle signal PJEN to jump one phase period. The FSLOW signal component of the output of adder 16 provides the direction of the phase jump, which depends on whether an overflow or underflow condition occurred. It should be noted that the adder 122 does not reset when an overflow or underflow condition occurs. This allows the residue of the previous cycle to be added to the present integral-plus-proportional terms via 16-bit register 124. This further reduces the overall jitter of the sample clock output 16 provided by the multi-phase clock generator 24 (Fig. 4) by eliminating truncation errors. Those familiar with the art will understand that, in the interests of efficiency and conserving integrated circuit die area, the PDM accumulator, second-order loop filter system 100 may be implemented utilizing an 8-bit ALU as its core, the ALU being time-multiplexed to perform the various arithmetic functions of the system 100. The control and bandwidth selection functions of the filter 100 may then be implemented utilizing conventional circuitry. Figs. 6A-6I combine to provide a schematic diagram of an embodiment of a PDM accumulator, second-order loop filter circuit 100 that utilizes a time-multiplexed ALU in accordance with the present invention. The core of the circuit 100 is provided by an 8-bit arithmetic logic unit (ALU) 200, shown in Fig. 6, which performs different arithmetic functions during different groups of its cycles. Brief reference to the Fig. 7 time slot allocation diagram will be helpful in understanding the time-multiplexed operation of the ALU. The upper waveform in Fig. 7 represents the 15.36 MHz system clock that drives the PDM accumulator, second-order loop filter circuit 100. The second waveform in Fig. 7 represents the 80 KHz sample clock output 16 of the stepped clock generator 24 (Fig. 4). The next four waveforms represent, in sequence, the time-multiplexed operation of the ALU 200 at four different bandwidth settings (1X, 2X, 4X, 8X) of the PDM accumulator operation performed by the ALU 200. The 1X, 2X, 4X and 8X bandwidth settings correspond to 3, 6, 12 and 24 PDM cycles, respectively. As shown in Fig. 7, during a one cycle group of the system clock, the ALU 200 performs the function of the integrator/accumulator 108 (Fig. 5), adding the current 16-bit phase error input signal, which is generated by ROMs 201a and 201b and then provided at the outputs of D flip-flops 202, 204, and the previously calculator integral term retrieved from four 8-bit integral term ledger registers 206, 208, 210, 212; the result of this integrator/accumulator operation is then stored to the four integral term ledger registers 206, 208, 210, 212, replacing the previous term. The ROMs 201a and 201b utilize a full word control code SGSEL1.SRC and SGSEL2.SRC, respectively, to generate the phase error input (the source listings for these two programs are provided as Appendix A at the end of the detailed description. During a second cycle group, the ALU 200 performs the function of the integrator-plus-proportional accumulator 110 (Fig. 1), adding the current 16-bit phase error input signal and the previously-calculated integral-plus-proportional term retrieved from two 8-bit integrator-plus-proportional ledger registers 214, 216; the result of this integral-plus-proportional accumulator operation is then stored to the integral-plus-proportional ledger registers 214, 216; replacing the previous integral-plus-proportional term. During a third cycle group, the ALU 200 performs the function of the PDM accumulator 120 (Fig. 1), adding the current integral-plus-proportional term retrieved from the integral-plus-proportional ledger registers 214, 216 and the previously-calculated PDM accumulator term stored in two 8-bit PDM accumulator ledger registers 218, 220; the result of this PDM accumulator operation is then stored to the PDM accumulator ledger registers 218, 220, replacing the previous PDM accumulator term. During the PDM accumulator operation, the overflow/underflow of the ALU is monitored by a PDM controller that comprises a ROM 222 that stores an overflow/underflow and direction code PLLODD.SRC, the source listing for which is provided as Appendix B at the end of this detailed description. This code provides a 2-bit output PSLOW and PJEN via stabilizing latch 224. The PJEN bit indicates whether a phase jump is to be taken by the associated multi-phase clock generator 24; the PSLOW bit indicates the direction of the phase jump. The ALU 200 is controlled by an ALU controller comprising a ROM 226 and associated D flip-flop 228. The ROM stores a control code PLLCTL.SRC, the source listing for which is provided as Appendix C at the end of this detailed description, and responds to a set of control setting inputs provided to ROM 226 via latch 230. The clocking and output enable cycles that synchronize the retrieval of information from and storage of information to the various ledger registers for the corresponding operations of the ALU 200 is controlled by a ledger controller that includes ROMS 232 and 234 that stores code PLLCKC.SRC the source listing for which are provided as Appendix D at the end of this detailed description. Both the ALU controller and the ledger controller are driven by a sequencer that performs time slot allocation for the various cycle groups that the ALU 200 is to perform. A sequencer ROM 236 stores a program code PLLTSA.SRC, the source listing for which is provided as Appendix E at the end of this detailed description, which is a time slot allocation program for the ALU. As stated above, Fig. 7 provides a timing diagram illustrating time slot allocation for the PDM accumulator, second-order loop filter system 100 shown in Figs. 6A-6I. Time slot allocation waveforms are shown for four bandwidth settings: 1X, 2X, 4X and 8X. Inspection of the time slot allocation waveforms will show that the 1X, 2X, 4X and 8X bandwidth settings provide 3, 6, 12 and 24 PDM cycles, respectively, per sample clock period. Each PDM cycle is illustrated as a positive step within the PDM cursor boundary. The time slot allocation signals shown in Fig. 7 were obtained utilizing a digital-to-analog converter at the outputs of a 4-bit D flip-flop, which was used to stabilize the monitored signals. The inputs of the D flip-flop were connected to pins 1-4 of controller ROM 226. The clock of the D flip-flop was connected to CLK input of component 228. As further shown in Fig. 7, there are 96 cycles of the system clock in one sample clock period (12.5 µsec. The source program provided as Appendix F at the end of this detailed description provides the details of these cycles. It should be understood that various alternatives to the embodiment of the invention described herein may be employed in practicing the invention. It is intended that the following claims define the scope of the invention and its structure and methods within the scope of these claims be covered thereby.

Claims: A digital phase locked loop (DPLL) for tracking changes in the frequency of an incoming analog signal, the DPLL having: (a) a multi-phase clock generator responsive to a control signal for selecting one of a plurality of phase-separated clock signals generated by the multi-phase clock generator as a sample clock output signal; (b) phase comparator means for generating a multi-bit phase error signal corresponding to the phase difference between the incoming analog signal and the sample clock output signal,characterized by: (c) proportional means (102) responsible to the multi-bit phase error signal for generating a current multi-bit proportional term; (d) integrator accumulator means (108) for accumulating the current proportional term to generate a current integral term; (e) summing means (110) responsive to the current proportional term and the current integral term for generating a current multi-bit integral-plus-proportional term; and (f) pulse density modulation (120) accumulator means for accumulating the current multi-bit integral-plus-proportional term to generate the control signal, the control signal comprising components indicative of both the number of phase jumps to be taken by the multi-phase clock generator and the direction of the phase jumps.

Inventors: GUINEA JESUS, WILSON HOWARD, WONG HEE, GUINEA, JESUS, WILSON, HOWARD, WONG, HEE
Applicants: NAT SEMICONDUCTOR CORP, NATIONAL SEMICONDUCTOR CORPORATION
Application year: 1991
Main classification: H04L7
IPC codes: H03L7, H04L7