Far-end echo canceller

Abstract: A far-end echo path may be simulated at a terminal which utilizes QAM by processing the in-phase and quadrature data symbols by a pulse shaping filter (51) whose transfer function may be a raised cosine function. The in-phase and quadrature data symbols are then multiplied by cos(ω c +ω o )t and sin(ω c +ω o )t , respectively, where ω c is the carrier frequency and ω o is the adaptively determined offset frequency of the far-end echo signal. The output signals of the multipliers are then summed (94) and the result is processed by a single transversal filter (100) whose transfer function is determined adaptively. The output of this filter is a simulated far-end echo signal.

Description: The present invention relates to a data transmission system including an echo canceller. More particularly, the present invention relates to a far-end echo canceller and a circuit for simulating a far-end echo path for use in a communication system in which the far-end echo contains a frequency offset. It also relates to a method for simulating a far-end echo path in a communication system. In a typical full duplex data communication system, a local terminal simultaneously transmits data to and receives data from a remote terminal via a communication channel which is formed at least in part by a two-wire telephone line. A hybrid coupler is located at each end of the channel for the purpose of isolating the incoming and outgoing signals. Because the hybrid couplers operate imperfectly, the outgoing signal of the local terminal may be partially reflected at the local hybrid coupler in the form of a near-end echo and partially reflected at the remote hybrid coupler in the form of a far-end echo. Both echoes corrupt the incoming signal of the local terminal. The near-end echo is generally identical in carrier frequency to the local outgoing signal. It can therefore be eliminated easily by a conventional near-end echo canceller. The near-end echo canceller comprises an adaptive linear transversal filter that simulates the transfer function of the near-end echo path that produced the near-end echo. The outgoing signal is processed by the near-end echo canceller to generate a simulated near-end echo signal. The simulated near-end echo signal is then subtracted from the incoming signal which is corrupted with the near-end echo. Correcting the far-end echo is not as simple because the far-end echo may have a continually shifting phase, for example, in the form of a frequency offset, relative to the original outgoing signal. This may be caused by small differences between the carrier frequency of the outgoing signal and the carrier frequency of the far-end echo signal that occur when circuits used to step up and step down the signal carrier frequency are not perfectly matched. This is the case when the communications channel between the local data terminal and the remote data terminal includes a satellite hop. Because of the non-linearity in the far-end echo transfer function as represented by the changing phase, a far-end echo canceller in the form of a linear transversal filter is unable to adequately simulate the far-end echo transfer function (see, e.g., Ling, U.S. Patent 4,813,073). Thus, conventional far-end echo cancelers are quite complex. For example, in the case where Quadrature Amplitude Modulation (QAM) is utilized to modulate the data symbols onto a carrier frequency of ω c /2π, a conventional far-end echo canceller for simulating a far-end echo path comprises four transversal filters, followed by two summers, followed by two multipliers, followed by another summer. In Electronics Letters, vol. 22, No. 4, 13th February 1986, pages 199,200, Stevenage GB, A. Fagan et al.: "In-Band Least-Squares Echo Canceller" a transmitter is revealed. The data to be transmitted are processed by a QAM modulator and pulse-shaping filters. The transmitter also comprises a near-end echo canceller and a far-end echo canceller for simulating the near-end and the far-end echo path both being connected to the output of the QAM modulator. The far-end canceller has complex taps, a discrete phase-locked loop (DPLL) and a summer to track phase jitter and frequency offsets. As input for the data symbols the far-end echo canceller uses QAM modulated signals. Furthermore, in 1989 Internat. Conference on Acoustics, Speech Processing 2 Digital Signal Processing S2D, vol. 2, 23.05. - 26.05.1989, pages 1384 to 1387, A. Alvarez: 'Echo Canceller Design and Implementation of a CCITT V.32 Modem : Software Solution' a circuit for simulating a far-end echo path in a communication system is revealed. It comprises a multiplier, means for obtaining the real part of the multiplier output signal and a single transversal filter having a transfer function for filtering the real part of the multiplier output signal to output a simulated far-end signal. Also a near-end echo canceller is described containing a pulse-shaping filter for obtaining the near-end echo cancellation. The disadvantage of these known far-end echo cancellers is that they are very complex. Accordingly, it is an object of the present invention to provide a far-end echo canceller for use in a system where the far-end echo signal system has a frequency offset, which echo canceller is simpler in construction than the above-described conventional echo cancellers. Furthermore, it is an object of the present invention to provide a far-end echo canceller which is of simple construction for a communication system which utilizes QAM. It is also an object of the invention to provide a simple and effective method for simulating a far-end echo path in a cmmunication system which uses QAM. One of the objects is met by a method for simulating a far-end echo path in a communication system which utilizes Quadrature Amplitude Modulation wherein a simulated far-end echo signal is used to cancel far end echos, said method comprising the steps of after in-phase and quadrature data symbols have been filtered by a pulse shaping filter means, multiplying said filtered in-phase and quadrature data symbols by cos(ω c +ω o )t and sin(ω c +ω o )t, respectively, where ω c is a carrier frequency and ω o is a variable offset frequency of said far-end echo path, to produce first and second multiplied signals, summing the first and second multiplied signals, and processing the summed signal with a single transversal filter with a transfer function having adaptively determined coefficients to obtain the simulated far-end echo signal. The other object is met by an echo canceller to correct for frequency offset of a far-end echo path in a communication system, which utilizes Quadrature Amplitude Modulation, said echo canceller comprising summing means, said echo canceller being characterized by multiplication means for respectively receiving filtered in-phase and quadrature data symbols after said in-phase and quadrature data symbols have been filtered by a pulse shaping filter means, and for multiplying said filtered in-phase and quadrature data symbols by cos(ω c +ω o )t and sin(ω c +ω o )t, respectively, where ω c is the carrier frequency and ω o is a variable offset frequency of a far-end echo path of said communication system, to produce first and second multiplied signals , said summing means being disposed for summing the first and second multiplied signals, and a single transversal filter for outputting a simulated far-end echo signal, said single transversal filter being in communication to the output of said summing means and having a single transfer signal input and a transfer function with adaptively determined coefficients. The last of the objects is met by a circuit for simulating a far-end echo path to correct for frequency offset of the far-end echo path in a communication system which uses Quadrature Amplitude Modulation, said circuit comprising a multiplier, means for obtaining the real part of the multiplier output signal, the output of said obtaining means being in communication with the input of a single transversal filter having a transfer function with adaptively determined coefficients for filtering the real part of the multiplier output signal to output a simulated far-end echo signal, said circuit being characterized by a pulse shaping filter for filtering data symbols and said multiplier being in communication with said pulse shaping filter for multiplying said filtered data symbols by a sinusoidal signal whose frequency is the sum of the carrier frequency and a variable offset frequency of said far-end echo path. In accordance with an illustrative embodiment of the present invention, a far-end echo path may be simulated at a terminal which utilizes QAM by processing the in-phase and quadrature data symbols by a pulse shaping filter whose transfer function is, for example, a raised cosine function. The in-phase and quadrature data symbol are then multiplied by cos(ω c +ω o )t and sin(ω c +ω o )t, respectively, where ω c is the carrier frequency utilized by the QAM and ω o is the offset frequency which is determined by a phase lock loop adaptively. The output signals from the multipliers are then summed using a summer and the summed signal is then processed by a transversal filter having a real transfer function which is adaptively determined. The output of this filter is a simulated far-end echo signal. Because the pulse shaping filter is typically incorporated into the transmitter, the echo canceller may be implemented using only two multipliers, one summer and one transversal filter. This is much simpler than the conventional far-end echo canceller described above. FIG 1 schematically illustrates a communication system including a far-end echo canceller. FIG 2 schematically illustrates a transmitter for use in the communication system of FIG 1. FIG 3 illustrates a conventional far-end echo canceller for use in the communication system of FIG 1. FIG 4 illustrates a far-end echo canceller for use in the system of FIG 1, in accordance with an embodiment of the present invention. FIG 5A and FIG 5B explain the mathematical relationship between the echo cancellers of FIG 3 and FIG 4. FIG 1 schematically illustrates a communication system 1. The system comprises a local terminal 10 and a remote terminal 10'. The local terminal 10 transmits data to and receives data from the remote terminal 10' via communications channel 12. The channel 12 is formed in part by a two-wire telephone line. However, portions of the channel 12 may be formed by other communications media such as a satellite hop. Data symbols to be transmitted from the terminal 10 arrive at a transmitter 14 via line 11. At the transmitter 14, the data symbols are modulated into a carrier using, for example, Quadrature Amplitude Modulation. The modulated carrier signal is then routed to a hybrid coupler 16 via line 17. The hybrid coupler 16 routes the modulated carrier signal to a two wire line 13 which forms part of the communications channel 12. Similarly, at the remote terminal 10', a transmitter 14' receives data symbols via line 11' and modulates the symbols onto a carrier. The modulated carrier is then delivered to a hybrid coupler 16' via line 17'. The hybrid coupler 16' then routes the modulated carrier signal onto a two-wire line 13' which forms part of the channel 12. Thus the two-wire lines 13 and 13' each carry both an outgoing modulated signal and an incoming modulated signal. At the local terminal 10, the hybrid 16 routes the incoming signal on the two-wire line 13 to the summers 18 and 19 and the receiver 22 via the line 27. Similarly, at the remote terminal 10', the hybrid 16' routes the incoming signal on the two-wire line 13' to the summers 18' and 19' and the receiver 22' via the line 27'. The summers 18, 19 and 18', 19' are utilized for echo cancellation in a manner described below. After echo cancellation, the receivers 22 and 22' demodulate the received modulated carrier signal to provide output data symbols on the lines 31 and 31', respectively. Because of imperfections such as impedance mismatches in the hybrids 16 and 16', echoes are generated in the system 1 of FIG 1. Thus, a portion of the outgoing modulated carrier on line 17 of the terminal 10 is actually transmitted by the hybrid 16 to the line 27 rather than to the two-wire line 13. Similarly, a portion of this same signal originating in the terminal 10 is reflected at the hybrid 16' back onto the two-wire line 13'. Thus, the incoming modulated signal on the line 27 in the terminal 10 comprises the incoming signal s(k) generated by the transmitter 14' in the terminal 10', a near-end echo e n (k) generated by the hybrid 16 imperfectly coupling the signal generated by the transmitter 14 to the two-wire line 13, and a far-end echo e f (k) resulting from imperfect coupling of the incoming signal on the two-wire line 13' by the hybrid 16' into the line 27'. To eliminate the near-end echo e n (k) and the far-end echo e f (k), the terminal 10 includes a near-end echo canceller 20 and a far-end echo canceller 30. The near-end echo canceller 20 generates a signal ê n (k) which is a simulation of the near-end echo signal e n (k). The far-end echo canceller 30 generates a signal ê f (k) which is a simulation of the far-end echo signal e f (k). The summer 18 is utilized to subtract ê f (k) from the signal on line 27 to produce a signal on line 26 y(k)=s(k) + e n (k) . The summer 19 is utilized to subtract ê n (k) from y(k) to produce the signal s(k) on line 28. It should be noted that the signal on line 27' of the remote terminal 10' is also corrupted by near-end and far-end echoes. To cancel these echoes the remote terminal 10' also includes a far-end echo canceller 30' which is identical to the far-end echo canceller 30 and a near-end echo canceller 20' which is identical to the near-end echo canceller 20. As indicated above, the near-end echo is generally identical in carrier frequency to the local outgoing signal. For this reason, the near-end echo canceller 20 comprises a linear transversal filter that simulates the transfer function of the near-end echo path. Cancelling the far-end echo is not as simple because the far-end echo has a frequency offset. FIG 2 illustrates the transmitter 14 in greater detail. The transmitter 14 utilizes Quadrature Amplitude Modulation (QAM). The transmitter 14 receives a sequence of symbols x(k) to be transmitted at the splitter 50. As used herein, k=0,1,2,... is a discrete variable corresponding to the times t=kT where T is the sampling interval. The splitter 50 separates the symbols x(k) into in-phase symbols a(k) and quadrature symbols b(k). The symbols a(k) and b(k) are then processed by pulse shaping filters 51 with the transfer function g(k). The filters g(k) are pulse shaping filters which are utilized to limit the frequency spectrum in the frequency domain and to reduce intersymbol interference in the time domain. Illustratively, the transfer function g(k) is a raised cosine function. The QAM modulator 52 comprises two multiplier 53a, 53b and one summer 54. The output of the modulator 52 is represented by the symbol x(k) where x(k)=a(k)cosω c t+b(k)sinω c t, and where ω c /2π is the carrier frequency. The symbol x(k) is transmitted to the pass-band filter 55 and then to the hybrid 16. In some cases the pulse shaping filters 51 may be located following the modulator 52 and before the pass-band filter 55. In addition, an analog-digital converter (not shown) may be included in the transmitter before the pass-band filter 55. A conventional circuit for simulating the far-end echo path is shown in FIG 3. This circuit comprises three stages, a first rotator stage 60, a set of crosscoupled transversal filters 70, and a post rotator stage 80. The rotator stage 60 receives the quadrature and in-phase data symbols a(k), b(k) and outputs the symbols a'(k), b'(k) . The rotator 60 comprises four multipliers 62 for multiplying a(k) by sinω c t and cosω c t and for multiplying b(k) by sinω c t and cosω c t. The rotator 60 also comprises two summers 64. The set of transversal filters 70 comprises four crosscoupled linear transveraal filters 71, 72, 73, 74 with transfer functions c(k), d(k), d(k), and c(k), respectively. The coefficients of the transfer functions are determined adaptively in a conventional manner. The outputs of the filters 71 and 73 are combined using the summer 75 and the outputs of the filters 72 and 74 are combined using the summer 76. The rotator 80 is utilized to rotate the signal at the offset frequency ω o /2π. The offset frequency is determined adaptively using a phase-locked loop circuit 81. The multiplier 82 multiplies the output of the summer 75 by cosω o t and the multiplier 83 multiplies the output of the summer 76 by sinω o t. The outputs of the multipliers 82 and 83 are added using the summer 84 to produce the simulated far-end echo signal ê f (k). The circuit of FIG 3 may be utilized to implement the far-end echo canceller 30 of FIG 1. However, this echo canceller is complex, comprising four transversal filters 71, 72, 73, 74, three summers 75, 76, 84, and two multipliers 82, 83. In accordance with the present invention, the circuit for simulating the far-end echo path can be simplified in comparison to the conventional circuit for simulating the far-end echo path illustrated in FIG 3. A circuit for simulating the far-end echo path in accordance with the present invention is illustrated in FIG 4. In FIG 4 the far-end echo path is simulated by applying the pulse-shaping filters 51 to the quadrature and in-phase data symbols a(k) and b(k). The symbols a(k) are then multiplied by cos(ω c +ω o )t using the multiplier 90 and the symbols b(k) are multiplied by sin(ω c +ω o )t using the multiplier 91. In this case ω c /2π is the carrier frequency and ω o /2π is the offset frequency which is adaptively determined using the phase lock loop circuit 92. The outputs of the multipliers 90 and 91 are summed using the summer 94 to generate a signal q(k). The signal q(k) is processed by a single linear transversal filter 100 with transfer function f(k). The coefficients of this linear transversal filter are determined adaptively using a Least Mean Square (LMS) algorithm. The output of the filter 100 is then estimated far-end echo signal ê f (k). It should be noted that the filters 51 are contained in the transmitter as shown in FIG 2. Thus, a far-end echo canceller (see element 30 of FIG 1) can be implemented using two multipliers 90, 91, one summer 94, and a single linear transversal filter 100. This is far simpler than the conventional far-end echo canceller illustrated in FIG 3. The operation of the filter 100 may be understood in greater detail below. The estimated far-end echo signal ê f (k) is determined by the formula where f i (k), i=0,1,...n-1 are coefficients of the linear transversal filter 100 and n is an integer indicating the number of such coefficients. The coefficients are determined according to the LMS algorithm as follows where µ is a step size and y(k) is the signal on line 26 of FIG 1. The mathematical relationship between the two ways to simulate the far-end echo path shown in FIG 3 and FIG 4 can be understood in connection with FIG 5A and 5B. In FIG 5A and 5B, double lined arrows represent complex quantities and single lined arrows represent real quantities. The far-end echo path simulator circuit of FIG 3 is redrawn in FIG 5A. In FIG 5A the rotator stage 60 of FIG 3 is represented by the complex modulator 110 so that A(k)=a(k)+jb(k), A'(k)=a'(k)+jb'(k) and A'(k)=A(k)e jωct . The transversal filter stage 70 of FIG 3 is represented in FIG 5A by the complex transversal filter 112 whose transfer function h(k)=c(k)+jd(k). Similarly, the rotator stage 80 of FIG 3 is represented by the complex modulator 114. To obtain the simulated far-end echo ê f (k), the real part of the output of the complex modulator 114 is obtained using the circuit 116 which corresponds to the summer 84 of FIG 3. The inventive far-end echo path simulation circuit is redrawn in FIG 5B. The real filters 51 of FIG 4 are represented in FIG 5B by the complex filter 212 with the transfer function g(k). The multipliers 90 and 91 are represented by the complex modulator 214 and summer 94 is represented by the circuit 216 which takes the real part of the output of the complex modulator 214 to produce the signal q(k). As in FIG 4, the signal q(k) is processed by the filter 100. The circuits of FIG 5A and 5B generate the same output signal ê f (k) when the filter 100 has a transfer function f(k) such that f(k) = Re[F a (k)e jωct ] where F a (k) is the analytic part of F(k) and h(k) = [g(k) e jωct *F(k)]. In short, a far-end echo canceller for a system which utilizes QAM and which has frequency offset has been disclosed. The inventive far-end echo canceller is far simpler than conventional far-end echo cancellers for use in QAM systems with frequency offset. Finally, the above-described embodiments of the invention are intended to be illustrative only. Numerous alternative embodiments may be devised by those skilled in the art without departing from the scope of the following claims.

Claims: A method for simulating a far-end echo path in a communication system (1) which utilizes Quadrature Amplitude Modulation wherein a simulated far-end echo signal is used to cancel far end echos, said method comprising the steps of after in-phase and quadrature data symbols have been filtered by a pulse shaping filter means (51), multiplying said filtered in-phase and quadrature data symbols by cos(ω c +ω o )t and sin(ω c +ω o )t, respectively, where ω c is a carrier frequency and ω o is a variable offset frequency of said far-end echo path, to produce first and second multiplied signals, summing the first and second multiplied signals, and processing the summed signal with a single transversal filter (100) with a transfer function having adaptively determined coefficients to obtain the simulated far-end echo signal. The method for simulating a far-end echo path in a comnunication system (1) of claim 1 wherein said coefficients of said transfer function of said transversal filter (100) for processing the summed signal are determined using an adaptive least mean square algorithm. An echo canceller to correct for frequency offset of a far-end echo path in a communication system (1) which utilizes Quadrature Amplitude Modulation, said echo canceller comprising summing means (94), said echo canceller being characterized by multiplication means (90,91) for respectively receiving filtered in-phase and quadrature data symbols after said in-phase and quadrature data symbols have been filtered by a pulse shaping filter means (51), and for multiplying said filtered in-phase and quadrature data symbols by cos(ω c +ω o )t and sin(ω c +ω o )t, respectively, where ω c is the carrier frequency and ω o is a variable offset frequency of a far-end echo path of said communication system (1), to produce first and second multiplied signals , said summing means (94) being disposed for summing the first and second multiplied signals, and a single transversal filter (100) for outputting a simulated far-end echo signal, said single transversal filter (100) being in communication to the output of said summing means (94) and having a single transfer signal input and a transfer function with adaptively determined coefficients. The echo canceller of claim 3 characterized in that it comprises said pulse shaping filter means (51). The echo canceller of claim 3 or 4 characterized in that said pulse shaping filter means (51) has a raised cosine transfer function. The echo canceller of claim 3 characterized in that it includes means for adaptively determining said offset frequency ω o by a phase locked loop (92). The echo canceller of claim 3 characterized in that said transversal filter (100) implements a transfer function having coefficients that are adaptively determined by using an adaptive least mean square algorithm. A circuit for simulating a far-end echo path to correct for frequency offset of the far-end echo path in a communication system (1) using Quadrature Amplitude Modulation said circuit comprising a multiplier (214), means (216) for obtaining the real part of the multiplier output signal, the output of said obtaining means (216) being communication with the the input of a single transversal filter (100) having a transfer function with adaptively determined coefficients for filtering the real part of the multiplier output signal to output a simulated far-end echo signal, said circuit being characterized by a pulse shaping filter (212) for filtering data symbols and said multiplier (214) being in communication with said pulse shaping filter (212) for multiplying said filtered data symbols by a sinusoidal signal whose frequency is the sum of the carrier frequency and a variable offset frequency of said far-end echo path.

Inventors: TZENG CHIN-PYNG JEREMY, TZENG, CHIN-PYNG JEREMY
Applicants: IND TECH RES INST, INDUSTRIAL TECHNOLOGY RESEARCH INSTITUTE
Application year: 1991
Main classification: H04B3
IPC codes: H04B3, H04B15