Arrangement for converting binary input signal into corresponding in-phase and quadrature phase signals

Abstract: In order to effectively reduce a memory size of each of two memories (60,62) provided in an arrangement for converting a binary input data into the corresponding in-phase and quadrature signals, a memory output controller(64) and a sequential logic (66) are provided. The memory output controller (64) includes two polarity control circuits (76,78) and two input data selectors (80,82). The two polarity control circuits (76, 78) are respectively coupled to the two memories (60,62), while the two input data selectors (80,82) are preceded by and coupled to both of the two polarity control circuits (76,78). Each of the two polarity control circuits (76,78) reverses the polarity of the output of the associated memory according to the output of the sequential logic. On the other hand, each of the two input data selectors (80,82) is arranged to selectively acquire the outputs of the two polarity control circuits (76,78) depending on the output of the sequential logic (66).

Description: The present invention relates to an arrangement for converting a binary input signal into corresponding in-phase and quadrature phase signals, and more specifically to an arrangement suitable for use in GMSK (Gaussian minimum shift keying) modulation. It is known in the art to use an arrangement as shown in Fig. 1 to convert a binary input signal into in-phase and quadrature baseband signals for GMSK quadrature modulation. Before discussing the known Fig. 1 arrangement in detail it is deemed advantageous to briefly discuss the principle underlying the same. It is known that a phase (t) of a GMSK modulating signal is represented by equation (1). wherein T: bit period; d i : i-th incoming data [0, 1]; a i : i-th symbol [+1, -1] (=1-2d i ); σ = Ln(2) /2πBT; B: 3 dB cut-off frequency of GMSK baseband filter; and Equation (1) can be approximated as equation (2). wherein M is an appropriate natural number. If M≥2, a phase error using the approximation equation (2) is very small (viz., within ±0.2°). Equation (2) can be rewritten using a given positive integer k indicating the number of data as follows. where T(k-1/2) < t < T(k + 1/2).  k (t) represents a phase of a symbol a k , viz., the phase during a time duration defined between T(k-1/2) and T(k+1/2). Designating the first and second terms of equation (3) by X k (t) and Y k respectively, the following equation is given  k (t) = X k (t) + Y k It is understood from equation (3) that X k (t) is determined by a k and N symbols preceding and following a k (viz., (N+1) symbols in total). That is to say, X k (t) is determined by d k and the M-bit preceding and following d k (viz., (M+1)-bit in total). On the other hand, Y k can be represented by Y k = Y k-1 + (π/2) a k-M-1 Accordingly, Y k is defined by the past status of a symbol and hence becomes uncertain depending on initial conditions. However, the GMSK modulation is in fact implemented by a phase difference between adjacent symbols and therefore, there is no need for considering the absolute value of each phase. This means that if Y k satisfies the difference equation (5), the initial value thereof can be set to an arbitrary one. Since a i takes an either value of +1 or -1, if the initial value is assumed 0, Y k assumes one of the four values as shown in the following. Y k = 0, ±π/2, π (radian) Let us introduce two 2-bit variables of state p k , q k and have Y k correspond to them as follows. Fig. 2 is a sketch showing the transitions of the state of Y k which can be obtained from equations (5) and (7). In Fig. 2, the values of Y k (0, +π/2, π and -π/2) are respectively indicated within circles. It is understood, from equations (3), (4) and Fig. 2, that  k (t) is determined by the following: (a) d k ; (b) M bits which precede d k ; (c) M bits which follow d k ; and (d) p k and q k defined by equation (7). On the other hand, the in-phase signal (I k (t)) and the quadrature signal (Q k (t)) are given by I k (t) = cos  k (t) Q k (t) = -sin  k (t) By defining discrete values of time as shown in equation (9), t L = T(k-1/2) + (T/N)L + T/2N wherein L is a discrete variable (L = 0, 1, ···, (N-1)), and N is a positive integer we obtain X k (t L ) is determined by a k-M , a k-M+1 , ···, a k , ···, a k+M-1 , a k+M , and L. Accordingly, each of  k (t L ), I k (t L ) and Q k (t L ) is determined by d k-M , d k-M+1 , ···, d k , ···, d k+M-1 , d k+M , p k , q k and L. It follows that the in-phase and quadrature signals (I k (t L ), Q k (t L )) can be obtained by binary input data d i using two ROMs which respectively store I k (t L ) = cos  k (t L ) Q k (t L ) = sin  k (t L ). In this case, the data within the ROMs are derived using an address determined by d k-M , d k-M+1 , ···, d k , ···, d k+M-1 , d k+M , p k , q k and L. The above-mentioned principle is utilized in configuring the known arrangement shown in Fig. 1 wherein it is assumed that M=2 and N=8. The Fig. 1 arrangement will be discussed with reference to Fig. 3. A shift register 20 receives the binary input data d i via an input terminal 22 in synchronism with a clock signal /A2 and shifts the data d i according to the clock signal /A2. Throughout the instant specification and claims, the symbol "/" preceding "A2" (for example) denotes an inverted "A2" and corresponds to a bar over "A2" in the drawing. The clock signal /A2 is derived from an output terminal Q2 of an octal counter 24 via an inverter 26. The shift register 20 outputs d k-2 , d k-1 , d k , d k+1 and d k+2 in this particular case (viz., M=2) from five shift stages, which are inputted to each of two ROMs 28 and 30 as the upper 5-bits (viz., A5, A6, A7, A8 and A9) of a 10-bit address signal (denoted by ADD-28, ADD-30). The ROM 28 pre-stores the data of cos  k (t L ) while the ROM 30 pre-stores the data of sin  k (t L ). The variables of state p k and q k are obtained by p k = p k-1 ⊕ q k-1 ⊕ d k-M-1 q k = /q k-1 wherein d k-M-1 equals d k-3 (M=2) and ⊕ represents a logical operation of exclusive-OR. A sequential logic 32 is an arrangement for implementing the logical operations given by equation (11) and which includes two exclusive-OR gates 34, 36 and two D-type flip-flops 38, 40. The sequential logic 32 is supplied with d k-2 and the timing clock /A2 and outputs the state variables p k and q k both of which are applied to the ROMs 28, 30 as address bits A3, A4. The operation of the sequential logic 32 will readily be understood and hence further description thereof will be omitted for brevity. The octal counter 24 receives, via an input terminal 28, a clock signal CLK whose timing chart is shown in a row labelled CLK in Fig. 3. The counter 24 produces three address bits A0, A1 and A2 at the output terminals Q0, Q1 and Q2, which correspond to the discrete-time variable L and which are applied to the ROMs 28 and 30 as lower 3-bit of the addresses (viz., ADD-28, ADD-30). Fig. 3 shows a timing chart of each of the above-mentioned signals. For the convenience of a better understanding, a logical equation pk = p k-1 ⊕ q k-1 ⊕ d k-3 is inserted in the drawing. Each of the ROMs 28, 30 output respectively the data according to the addresses ADD-28, ADD-30 applied thereto. Two digital-to-analog converters (DACs) 42, 44 are provided for converting the digital outputs of the ROMs 28, 30 into corresponding analog signals, respectively. The analog signals thus obtained are derived via output terminals 46, 48 to an external circuit (not shown), and are used to modulate a carrier signal as is well known in the art. However, the aforesaid known technique has encountered the problem in that each of the ROMs 28, 30 must have an undesirably large storage capacity. More specifically, the number of address bits for each of the ROMs 28, 30 is ten (10) in the Fig. 2 arrangement, and hence the number of words required for each of the ROMs 28, 30 reaches 1024. In the case where the word length of each of the ROMs 28, 30 is 8-bit, the total number of bits required for the ROMs 28, 30 amounts to 16k-bit. Viz.,: (1024-word) x (8-bit/word) x 2 = 16k-bit Accordingly, requiring such a large memory leads to undesirably large memory chip sizes. It is therefore highly desirable to decrease the memory size of each of the ROMs 28, 30 for effectively reducing memory manufacturing costs, etc. EP-A-0169 612 discloses an arrangement for generating a constant amplitude angle modulated carrier signal in response to data signals. In particular, a generator produces two selection signals, wherein the first selection signal is applied to a two-input multiplexer which is connected to sections of the first read memory. In response to each clock signal, only one of the two signal samples is applied to a DAC circuit in accordance with the value of the first selection signal. A sine inverter is connected between the multiplexer and the DAC circuit. A second selection signal is applied to a second sine inverter so as to selectively invert the signal of the signal samples selected in the multiplexer at each clock instant based on the value of the second selection signal. It is an object of the present invention to provide an arrangement which features an effective reduction of the memory size required for each of the ROM(s) provided therein. This object is solved with the features of claim 1. In brief, the above object is achieved by an arrangement which includes a memory output controller and a sequential logic. The arrangement is able to effectively reduce a memory size of each of two memories provided in an arrangement for converting binary input data into the corresponding in-phase and quadrature signals. The memory output controller includes two polarity control circuits and two input data selectors. The two polarity control circuits are respectively coupled to the two memories, while the two input data selectors are preceded by and coupled to both of the two polarity control circuits. Each of the two polarity control circuits reverses the polarity of the output of the associated memory according to the output of the sequential logic. On the other hand, each of the two input data selectors is arranged to selectively acquire the outputs of the two polarity control circuits depending on the output of the sequential logic. More specifically an aspect of the present invention comes in an arrangement for converting binary input data into corresponding in-phase and quadrature signals, comprising: a counter, the counter receiving a first clock signal and counting clock pulses of the first clock signal up to N (N is a natural number), the counter generating a plurality of counter outputs at a given time point; a shift register, the shift register including a plurality of shift stages from which shifted data are derived, the shift register acquiring the binary input data and shifting same in response to a second clock signal the frequency of which is 1/N of the first clock signal; a first memory section, the first memory section pre-storing a look-up table including cosine data and receiving an address signal which consists of first bit signals each derived from the shift stages and second bit signals each derived as the counter output from the counter, the first memory section producing an output defined by the address; a second memory section, the second memory section pre-storing a look-up table including sine data and receiving the address signal and producing an output defined by the address; a sequential logic, the sequential logic producing outputs in response to both the second clock signal and one of the shifted data of the shift register; and an output controller, the output controller being coupled to receive the outputs of the first and second memory sections, the output controller selectively reversing a polarity of each of the outputs of the first and second memory sections under control of the sequential logic, the output controller selectively steering the outputs of the first and second memory sections to first and second output terminals thereof under control of the sequential logic. The features and advantages of the present invention will become more clearly appreciated from the following description taken in conjunction with the accompanying drawings in which like elements are denoted by like reference numerals and in which: Fig. 1 is a block diagram showing a prior art arrangement for converting a binary input signal into corresponding two in-phase and quadrature signals, referred to in the opening paragraphs; Fig. 2 is a sketch for describing the operation of the Fig. 1 arrangement, referred to in the opening paragraphs; Fig. 3 is a timing chart depicting the operation of the Fig. 1 arrangement, referred to in the opening paragraphs; Fig. 4 is a block diagram showing the circuit arrangement which characterizes an embodiment of the present invention; and Fig. 5 is a timing chart depicting the operational characteristics of the Fig. 4 arrangement. A preferred embodiment of the present invention will be discussed with reference to Figs. 4 and 5. Before turning to Figs. 4 and 5 a principle underlying the present invention will be discussed. As previously mentioned, the in-phase (I k (t L )) and quadrature (Q k (t L )) signals are represented by: wherein Y k = 0, ±π/2 or π. Y k corresponds to the variables of state p k , q k as shown in equation (7). Equation (12) can be rewritten as follows: Accordingly, we obtain the following equation (14) from equations (7) and (13). It is understood from equation (14) that if a first ROM pre-stores the data of cos X k (t L ) while a second ROM pre-stores the data of sin X k (t L ), then the I and Q signals can be derived by inverting the outputs 6f the first and ROMs and/or by exchanging the outputs thereof, both depending on the variables of state p k , q k . The arrangement shown in Fig. 4 is based on the above-mentioned principle as expressed by equation (14), and includes a shift register 20', a counter 24', two ROMs 60 and 62, an output controller 64, a sequential 66, and two digital-to-analog converters (DACs) 42', 44'. The blocks 20', 24', 42' and 44' are respectively identical with the blocks 20, 24, 42 and 44 in Fig. 1 and hence will not be discussed in detail for brevity. It should be noted that the Fig. 4 arrangement is configured under the assumption of M=2 and N=8. The ROMs 60 and 62 respectively pre-store the data of cos X k (t L ) and sin X k (t L ). The sequential logic 66 comprises two D-type flip-flops 68, 70 and two exclusive-OR gates 72, 74. As shown, the shifted input data d k-2 is applied to one input terminal of the exclusive-OR gate 72, while the clock signal /A2 is applied to a clock terminal of each of the flip-flops 68, 70. The output of the exclusive-OR gate 72 is defined by "p k ⊕q k ⊕ d k-2 ", while one of the inputs to the exclusive-OR gate 74 is defined by "p k-1 ⊕ q k-1 ⊕ d k-3 ". The output controller 64 includes two output polarity controllers 76, 78 and two input data selectors 80, 82. The polarity controller 76 is supplied with the output of the exclusive-OR gate 74 (viz., p k ⊕q k ) and reverses the output of the ROM 60 in the event that p k ⊕ qk 1. On the other hand, the polarity controller 78 reverses the output of the ROM 62 in response to the variable of state /p k which assumes a logic 1 (viz., in the event that p k = 0). The polarity controller 76 includes a plurality of exclusive-OR gates arranged in parallel, each of which is supplied with the output of the ROM 60 at one input terminal thereof and each of which is supplied with the above-mentioned p k ⊕ q k . Similarly, the polarity controller 78 includes a plurality of exclusive-OR gates arranged in parallel, each of which is supplied with the output of the ROM 62 at one input terminal thereof and each of which is supplied with the variable of state /p k . The number of the exclusive-OR gates of each of the polarity controllers 76, 78 is equal to the output lines (eight for example) of the associated ROM. The input data selectors 80, 82 respectively select the outputs of the controllers 76, 78 in the event that the variable of state q k assumes a logic 0. Otherwise, the selectors 80, 82 respectively select the outputs of the controllers 78, 76. It should be noted that each of the ROMs 60, 62 is provided with eight address lines instead of ten lines as in the case of the prior art referred to in connection with the Fig. 2. This means that the memory capacity can be reduced to one-fourth as compared with the above-mentioned prior art. By way of example, in the event that M=2, N=8 and the word length is 8-bit, the number of words required is markedly reduced to 256 words from 1024 words in the case of the above-mentioned prior art. Accordingly, the number of bits required by each of the ROMs 60, 62 is 256 words x 8 bits/word x 2 = 4k bits In the above description, two separate ROMs 60 and 62 are provided. However, it is within the scope of the present invention to prepare a single ROM which includes two memory sections for respectively pre-storing the data of cos X k (t L ) and sin X k (t L ). Further, each of the ROMs 60 and 62 can be replaced with a random access memory (RAM), in the case of which the data (cos X k (t L ), sin X k (t L )) should be transferred to the associated RAM prior to the operation. Still further, the counter 24' is not limited to a counter which counts up to a multiple of two. The counter 24' may take the form which counts up an appropriate natural number (N), in which case the shift register 20' operates in synchronism of a clock signal whose frequency is 1/N. Fig. 5 is a timing chart illustrating the timing of the signals shown in Fig. 4.

Claims: An arrangement for converting binary input data into corresponding in-phase and quadrature signals, comprising: a) a counter (24') which receives a first clock signal and counts clock pulses of said first clock signal up to N (N is a natural number), said counter generating at a given time point a plurality of counter outputs which includes a second clock signal the frequency of which is 1/N of said first clock signal; b) a shift register (20') which includes a plurality of shift stages from which shifted data are derived, said shift register (20') being coupled to said counter (24') so as to acquire the binary input data and shifting same in response to said second clock signal from said counter (24'); c) a first memory section (60) having a look-up table including cosine data pre-stored therein, said first memory section being coupled to said shift register (20') and said counter (24') so as to receive an address signal which consists of first bit signals received from the shift stages and second bit signals received as the counter output from said counter, said first memory section producing an output defined by the address; d) a second memory section (62) having a look-up table including sine data pre-stored therein, said second memory section (62) being coupled to said shift register (20') and said counter (24') so as to receive said address signal and produce an output defined by said address; e) a sequential logic (66) coupled to said counter (24') and said shift register (20'), said sequential logic producing outputs in response to both said second clock signal and one of the shifted data of said shift register; and f) an output controller (64); characterized in that g) said output controller (64) includes g1) first and second output terminals; g2) first and second memory output polarity controllers (76, 78), respectively coupled to receive the outputs of said first and second memory sections (60 and 62, respectively), for selectively reversing a polarity of each of the outputs of said first and second memory sections under control of said sequential logic; and g3) first and second input data selectors (80, 82) for selectively steering the outputs of said first and second memory sections to first and second output terminals under control of said sequential logic, each of the first and second input data selectors being coupled to said first and second memory output polarity controllers and to said first and second output terminals. An arrangement as claimed in claim 1, wherein said sequential logic (66) is a circuit which performs the following logical operations: p k = p k-1 ⊕ q k-1 ⊕ d k-M-1 wherein ⊕ is the exclusive-OR operator; q k = /q k-1 wherein / is the logical inversion operator; p k , q k are variables of state at a time point k; k is a time point defined by said second clock signal; M = (P-1)/2 wherein P is the number of the shift stages; / implies division; and d k-M-1 are binary input data derived from said shift register (20') at a time point (k-M-1), (p k ⊕ q k ) being applied to said first memory output polarity controller (76), /p k being applied to said second memory output polarity controller (78), and q k being applied to said first and second input data selectors (80, 82). An arrangement as claimed in claim 1 or 2, wherein said sequential logic (66) includes: a first exclusive-OR gate (72) having first and second input terminals and receiving one of the outputs of said shift register at said first input terminal; a first D-type flip-flop (68) coupled to receive an output of said first exclusive-OR gate (72), said first D-type flip-flop (68) applying one ( Q ) of two outputs thereof to said second output polarity controller (78); a second exclusive-OR gate (74) having first and second input terminals and receiving the other output (Q) of said two outputs of said first D-type flip-flop (68) at the first input terminal thereof, and applying the output thereof to said first output polarity controller (76) and to the other input terminal of said first exclusive-OR gate (72); and a second D-type flip-flop (70) coupled to receive one of two outputs thereof and applying the other of the two outputs thereof to the second input terminal of said second exclusive-OR gate (74) and to said first and second input data selectors (80, 82). An arrangement as claimed in any of claims 1 to 3, wherein said first and second memory sections (60, 62) are included in a single memory unit.

Inventors: ICHIHARA MASAKI, ICHIHARA, MASAKI, ICHIHARA, MASAKI, C/O NEC CORPORATION
Applicants: NIPPON ELECTRIC CO, NEC CORPORATION
Application year: 1991
Main classification: H04L27
IPC codes: H04L27, H04L25, H03H17