Successive approximation A/D conversion method and apparatus

Abstract: An analog-to-digital converter (ADC) includes successive approximation control logic for generating ten-bit binary numbers, a digital-to-analog converter (DAC) (205) having a resistor string (301) and a weighted-capacitor array (303) for converting the ten-bit binary output of the control logic to a known analog voltage, and an analog comparator (207) for comparing the output of the DAC (205) to a reference voltage provided via a tap (305) to the mid-point of the DAC resistor string (301). The unknown analog voltage input to the ADC and the reference voltage are provided to the capacitor array (303) to precharge the array (303) to a voltage equal to the reference voltage minus the unknown analog voltage. The output of the DAC (205) is equal to the known analog voltage plus the reference voltage minus the unknown analog voltage. The known analog voltage is changed in successive conversion steps such that the output of the DAC (205) converges towards the reference voltage value. High accuracy conversion is achieved over a wide voltage range.

Description: This invention relates to a method for converting an unknown analog voltage having a value between a first and a second supply voltage to a binary signal. The invention also relates to analog-to-digital conversion apparatus. A method and apparatus for analog-to-digital conversion are known from the article "High Resolution A/D Conversion in MOS/LSI" by B. Fotouhi and D A Hodges, IEEE Journal of Solid-State Circuits, vol. SC-11,No. 6, December 1976, pages 136-142. A schematic diagram of the known ten-bit successive-approximation analog-to-digital converter (ADC) is shown in Figures 1A and 1B. The ADC includes three main elements: a control logic circuit 101, a digital-to-analog converter (DAC) 103, and an analog comparator 105. DAC 103 includes a resistor string comprised of resistors R101 through R116 for converting a four-bit binary value to an analog value and a weighted-capacitor array comprising capacitors C101 through C107 for converting a six-bit binary value to an analog value. Switches S101 through S125 are controlled such that any ten-bit binary value can be converted into an analog value, the four most significant bits of the ten-bit binary value being converted by the resistor string and the remaining six bits being converted by the capacitor array. The converter operation is as follows. With switch S118 open, switch S126 closed, switches S120 through S125 placed in position "2" and switch S119 placed in position "3" the bottom plates of capacitors C101 through C107 are connected to sample and hold the unknown analog voltage (VIN) to be converted. After opening switch S126, a successive-approximation search among the resistor string taps is conducted to find the voltage segment within which the stored unknown analog voltage lies. During this search the converter successively divides in half the voltage range in which the comparator has placed the unknown analog voltage. Then, with switches S101 through S119 set to connect the ends of the resistor which defines this segment to buses 118 and 119, the capacitor bottom plates are switched in a successive-approximation sequence until the voltage potential of the top plate of the capacitor array, i.e., the voltage level provided to comparator 105, is equal to ground potential. The following example is provided to aid in the understanding of the operation of the converter. Assume that the VREF equals 16 volts and the unknown analog voltage VIN equals 7.8 volts. With VREF equal to 16 volts the voltage drop across each resistor R101 through R116 is 1 volt. Prior to initiation of the successive approximation search, capacitors C101 through C107 are charged to -7.8 volts (top plates relative to bottom plates) following the procedure set forth in the preceding paragraph. The steps in the successive approximation search are as follows, each step taking one clock cycle. The search begins with switches S120 through S125 placed in their "2" positions, switch S108 closed and switch S119 set to position 2. Thus the voltage potential of the bottom plates of the capacitor array is set to 8 volts (the analog equivalent to binary value 1000000000), and the voltage potential of the top plates, which is provided to the minus input of comparator 105, is 0.2 volts. The comparator output indicates that VIN is less than 8 volts so during the next search step switch S108 is opened and switch S104 is closed providing a voltage potential of 4 volts to the capacitor bottom plates. The top plate voltage is now -3.8 volts. The comparator output indicates that VIN is greater than 4 volts. During the third search step switch S104 is open and switch S106 is closed providing 6 volts to the bottom plates of the capacitor array and -1.8 volts to the top plates. The fourth search step provides 7 volts to the bottom plates of the capacitors by closing switch S107 and setting switch S119 to position "1". The top plate voltage will be -0.8 volts. Thus, after four clock cycles it is determined that the unknown analog voltage VIN lies between 7 and 8 volts. At the onset of the search among capacitors C101 through C107, switches S107 and S108 are closed, switches S118 and S119 are positioned to provide 7 volts to bus 119 and 8 volts to bus 118, switches S120 through S124 are set to their "2" positions and switch S125 is set to position "1". After redistribution of the charge among the capacitors, the potential of the top plate of the capacitors will be -0.3 volts. For the sixth search step switches S125 and S124 are set to their "1" position. The potential of the top plate of the capacitor array will equalize at -0.05 volts. A total of ten search steps will be conducted as shown in the table provided below. The voltages shown in steps 7 through 10 have been rounded off at the hundredths decimal place. At the conclusion of the search the binary equivalent of VIN will be determined to be 0111110011, the last value to produce a "yes" answer to the comparison. Step Comparison Answer Binary Value 1 8.00V-VIN <= 0V No 1000000000 2 4.00V-VIN <= 0V Yes 0100000000 3 6.00V-VIN <= 0V Yes 0110000000 4 7.00V-VIN <= 0V Yes 0111000000 5 7.50V-VIN <= 0V Yes 0111100000 6 7.75V-VIN <= 0V Yes 0111110000 7 7.88V-VIN <= 0V No 0111111000 8 7.81V-VIN <= 0V No 0111110100 9 7.78V-VIN <= 0V Yes 0111110010 10 7.80V-VIN <= 0V Yes 0111110011 During the sample period the capacitor array is charged to -VIN (top plate voltage relative to bottom plate). The control logic begins the successive approximation sequence by setting the bottom plates of capacitors C101 through C107 at a voltage equal to VREF/2, the voltage at the midpoint of the resistor string. The voltage potential of the top plate of the capacitor array will therefore equal (VREF/2) - VIN during the onset of the successive approximation sequence. If the potential of VIN is between zero and VREF then the voltage swing on the top plate of the capacitor array is -VREF/2 to VREF/2. Thus, the voltage on the top plate of the capacitor array will be outside of voltage range defined by the supply rails for VREF between 0 and Vdd causing parasitic diodes to turn on which will remove charge from the capacitor array. A disadvantage associated with the known analog-to-digital converter is that to achieve high accuracy, the input voltage range is restricted, to around one half of the supply span. To achieve higher voltage ranges, converter accuracy must be sacrificed. An analog-to-digital converter is also known from EP-A-0 298 493 which seeks to achieve high accuracy conversion over a wide input signal voltage range. However, the analog-to-digital converter of this document is disadvantageous since the input voltage to the comparator thereof does not remain within the rail to rail voltage of the converter. US-A-4,908,620 and US-A-4,831,381 each disclose analog-to-digital conversion apparatus in which, during the successive approximation step, the first comparison is carried out between the input voltage and an initial value of the known voltage which is generally the middle value of the allowable input voltage range. However, the apparatus disclosed in these further documents nevertheless exhibits disadvantages particularly insofar as the manner in which attenuation of an input signal may arise. For example, in US-A-4,831,381, the attenuation of the voltage input to the inverting amplifier in this manner when low-amplitude signals are being handled by the converter proves problematic since the level of attenuation adversely affects the accuracy with which such signals can be processed within the converter. It is an object of the present invention to provide a method and apparatus for analog-to-digital conversion whereby high accuracy conversion may be achieved over a wide input signal voltage range. Therefore, according to one aspect of the present invention, there is provided a charge redistribution analog-to-digital conversion method for converting an unknown analog voltage having a value between a first and second supply voltage to a binary signal by means of a capacitor array formed of a plurality of capacitors each having a first and second plates, the first plates being connected to comparison means which is supplied by way of a supply voltage span, comprising the steps of: (a) generating a reference voltage which is dependent upon the values of said first and second supply voltages by dividing said first and second supply voltages such that the reference voltage has a value equal to one half the sum of said first and second supply voltages and supplying said reference voltage to said first plates of the capacitors; (b) providing said unknown analog voltage to the second plates of the capacitors for subtracting said unknown analog voltage from said reference voltage; (c) auto-zeroing the DC voltage of the comparison means (207) to a value equal to half the sum of the voltages (Vdd, Vss) forming the said supply span (d) simultaneously isolating said first and second plates from said reference voltage and said unknown analog voltage respectively; (e) providing a known voltage initially equal to one half the sum of said first and second supply voltages and generating a voltage equal to the sum of the output of said subtracting step and said known voltage; (f) comparing the voltage determined by the step (e) with said reference voltage; (g) generating a new value for said known voltage dependent upon the result of the comparison; and (h) repeating steps (e) through (g) until the voltage generated in step (e) converges to said reference voltage. According to another aspect of the present invention, there is provided a charge redistribution analog-to-digital converter for converting an unknown analog voltage to a binary signal by a method according to Claim 1, including a capacitor array formed of a plurality of capacitors each having a first and second plate, the first plate being connected to comparison means which is supplied by way of a supply voltage span, reference voltage generating means adapted to generate a reference voltage which is dependent upon the value of said first and second supply voltages by dividing said first and second supply voltages such that the reference voltage is equal to one half the sum of said first and second supply voltages, and which reference voltage generating means is connected to supply said reference voltage to said first plates of said capacitor array, said capacitor array forming add/subtract means adapted to receive said unknown analog voltage at the second plates thereof and to subtract said unknown analog voltage from the said reference voltage to provide a difference voltage auto-zeroing means arranged for auto-zeroing the DC voltage of the comparison means (207) to a value equal to half the sum of the voltages (Vdd, Vss) forming the said supply span, and including switching means controlled to simultaneously isolate the first and second plates from the reference voltage and said unknown analog voltage respectively, and further including voltage generating means adapted to generate a voltage equal to the sum of the output of the add/subtract means and a known analog voltage initially equal to one half the sum of said first and second supply voltages, wherein said comparison means is arranged to compare the output of said voltage generator means with said reference voltage, and control means are provided for controlling the repeated generation of said voltage equal to the sum of the output of the add/subtract means and a known voltage until the generated voltage converges to said reference voltage. One embodiment of the present invention will now be described by way of example, with reference to the accompanying drawings, in which:- Figures 1A and 1B are a schematic diagram of a prior art ten-bit successive-approximation analog-to-digital converter; Figures 2A and 2B are a top view schematic diagram of an analog-to-digital converter constructed in accordance with the present invention; Figures 3A through 3C are a schematic diagram of the digital-to-analog converter shown in Figure 2. Figures 4A and 4B are a schematic diagram of the analog comparator shown in Figure 2; Figure 5 is a schematic diagram of the differential amplifiers shown in Figure 4; Figures 6A through 6C are a schematic diagram of the comparator bias circuit shown in Figure 4; and Figure 7 is a schematic diagram of the delay circuit shown in Figure 4. Referring now to Figures 2A and 2B, there is seen a top view schematic diagram of a ten-bit successive-approximation analog-to-digital converter constructed in accordance with the present invention. The ADC illustrated in Figure 2 includes an input 201 for receiving an unknown analog signal VIN to be converted to a digital value, a control logic circuit 203, a digital-to-analog converter (DAC) 205, an analog comparator 207, and a set of output terminals 211 through 220 for providing the binary equivalent of input signal VIN. As will be explained in more detail below, DAC 205 and comparator 207 include improvements which distinguish the present invention from prior art analog-to-digital converters. Control logic circuit 203 generates the ten-bit binary output, B0 - B9, which is provided to ADC output terminals 211 through 220. Control logic circuit 203 also includes decoding logic for converting signals B9 - B6, the four most significant bits of the binary output, into control signals C1 - C17. Each one of the sixteen possible four-bit combinations of B9 - B6 enables a different set of adjacent control signals as shown in table 2. B9-B6 Control Signals Set HIGH B9-B6 Control Signals Set HIGH 0000 C1, C2 1000 C9, C10 0001 C2, C3 1000 C10, C11 0010 C3, C4 1010 C11, C12 0011 C4, C5 1011 C12, C13 0100 C5, C6 1100 C13, C14 0101 C6, C7 1101 C14, C15 0110 C7, C8 1110 C15, C16 0111 C8, C9 1111 C16, C17 DAC 205 is connected to receive binary output signals B0 - B5, control signals C1 - C17, and a sample signal SMPLB from circuit 203, reference voltages VREFM and VREFP from external voltage sources (not shown), and input analog signal VIN from input 201. The DAC provides two analog outputs, VR1 and DOUT, to the plus and minus inputs, INP and INM, respectively, of comparator 207. The comparator generates complementary outputs OUTP and OUTM, providing output OUTM through a multiplexer 223 to control logic circuit 203. Multiplexer 223 is provided for testing purposes. DAC 205 and comparator 207 will be discussed in greater detail below. Figures 3A through 3C are a schematic diagram of DAC 205. DAC 205 includes a resistor-string 301 for converting the four most significant bits of digital data received from circuit 203, and a capacitor array 303 for converting the six least significant bits of digital data received from circuit 203. Resistor string 301 includes sixteen resistors labeled R301 through R316 connected in series between reference voltage inputs VREFM and VREFP, each resistor having the same ohmic value, 750 ohms. Switches S301 through S317 connect reference voltage VREFM, the nodes between the resistors and reference voltage VREFP to one of two buses, labeled X and Y, the odd-numbered switches being connected to bus X and the even-numbered switches connected to bus Y. Switches S301 through S317 operate in response to control signals C1 through C17, respectively, provided by the control logic. The capacitor array includes seven capacitors, labeled C301A, C301B, C302, C304, C308, C316 and C332. The capacitance of these resistors is shown to be 0.2pf, 0.4pf, 0.8pf, 1.6pf, 3.2pf, 6.4pf, and 12.8pf, respectively. The top plates of all of these capacitors are connected to a common node, while the bottom plates of C301B, C302, C304, C308, C316 and C332 are connected to switches S322 through S327, respectively. Switches S322 through S327 permit connection of the bottom plates of their respective capacitors to either one of two buses, labeled A and B. The bottom plate of capacitor C301A is connected directly to bus B. Switches S322 through S327 are controlled by signals B0 through B5. Between resistor string buses X and Y and capacitor array buses A and B are two switches S318 and S319. Switch S318 selectively couples one of buses X or Y to bus A and switch S319 connects one of buses X or Y to bus B. Control signal POL operates switches S318 and S319. A tap 305 is located at the mid-point of resistor string 301. This tap from the node between resistors R308 and R309 provides a voltage, hereinafter referred to as VR1, equal to (VREFP + VREFM)/2. VR1 is provided as one output of the DAC and is also provided to the top plate of capacitor array 303 through switch S320. Another switch, S321 provides the unknown analog voltage VIN to bus B. Switches S320 and S321 are responsive to a signal SMPLB received from the control circuit to provide their input signals to the top plate of capacitor array 303 and bus B, respectively. A signal CONV, which is generated by comparator 207, is provided to switches S318, S319 and S322 through S327. All of the DAC switches discussed above are implemented as CMOS transmission gates. Switches S318, S319 and S322 through S327 are single-pole-double-throw switches. The operation of DAC 205 in concert with the control logic is as follows. During an initial sample period, the CONV signal is set low positioning switches S322 through S327 to connect the bottom plates of the capacitor array to bus B and opening switches S318 and S319 completely to disconnect buses X and Y from buses A and B. The sample signal SMPLB is also set low closing switches S320 and S321 to provide the unknown analog voltage VIN to bus B and reference voltage VR1 to the top plate of the capacitor array. The capacitor array is thereafter charged to store a voltage equal to VR1 - VIN. The sample period ends with signal SMPLB being set HIGH opening switches S320 and S321 to isolate the capacitor array from voltage signals VR1 and VIN. After auto-zeroing of the analog comparator, which will be discussed below, a successive-approximation search is conducted until the voltage potential of the top plate of capacitor array 303 is equal to VR1. The operation of control logic circuit 203 in concert with DAC 205 to convert ten-bit binary data into analog data through successive approximation proceeds in a generally similar manner to that described with reference to Fig. 1, except that successive approximations are continued until the top plate capacitor voltage has substantially converged to the value VR1. During the sample period the capacitor array is charged to VR1 - VIN. The control logic begins the successive approximation sequence by setting the bottom plates of capacitors C1 through C7 at a voltage equal to VR1, i.e., the voltage at the midpoint of the resistor string. The voltage potential of the top plate of the capacitor array will therefore equal (2 x VR1) - VIN during the onset of the successive approximation sequence. If the potential of VIN is restricted to be between VREFM and VREFP then the voltage swing on the top plate of the capacitor array is VREFP to VREFM, the supply voltage range for resistor string 301. A schematic diagram of analog comparator 207 of Figure 2 is shown in Figures 4A and 4B. Three fully differential amplifier stages with feedback switches and input capacitors are shown. The first stage is connected to the comparator plus and minus inputs, INP and INM. Input INP is provided through parallel-connected input capacitors C401 and C402 to the plus input of a differential amplifier 410. Similarly, input INM is provided through parallel-connected input capacitors C404 and C405 to the minus input of differential amplifier 410. Inputs INP and INM are also connected to voltage source VSS through capacitors C403 and C406, respectively, which represent the parasitic component of the parallel-connected input capacitors. Stage 1 also includes a feedback switch T411 connected between the minus output and plus input of amplifier 410 and a switch T412 connected between the plus output and minus input of the first stage amplifier. Both switches operate in response to a signal AZ1 generated by a delay circuit 409. Stages 2 and 3 are constructed similarly to stage 1. Stage 2 includes a differential amplifier 430 having a plus input connected to receive the minus output of amplifier 410 through a coupling capacitor C428 and a minus input connected to receive the plus output of amplifier 410 through a coupling capacitor C429. Switches T431 and T432 are provided between the minus output and plus input and the plus output and minus input, respectively, of amplifier 430. Switches T431 and T432 operate in response to a signal AZ2 generated by delay circuit 420. Stage 3 includes a differential amplifier 450 having a plus input connected to receive the minus output of the second stage amplifier through a coupling capacitor C448 and a minus input connected to receive the plus output of amplifier 430 through a coupling capacitor C449. Switches T451 and T452 are provided between the minus output and plus input and the plus output and minus input, respectively, of amplifier 450. Switches T451 and T452 are controlled by a signal AZ3 generated by delay circuit 440. The third stage is followed by a differential latch 460. In the preferred embodiment of the invention capacitors C401, C403, C404 and C406 are 0.63 pF capacitors and capacitors C402 and C405 are 0.37 pF capacitors. Capacitors C401 through C406 are poly/metal 1/metal 2 sandwich capacitors to reduce leakage current. Interstage coupling capacitors C428, C429, C448 and C449 are poly/diffussion type capacitors, each having a capacitance of 0.57 pF. Each comparator stage has a gain of about ten and generates an output signal which is proportional to the difference between the two input signals. The overall gain of the comparator (to the latch input) is about eight hundred as some attenuation occurs through interstage coupling. Transistor switches T411, T412, T431, T432, T451 and T452 are used to configure each comparator stage in a unity gain configuration during auto-zero mode. Latch 460 is connected to receive as inputs the plus and minus outputs of differential amplifier 450 and functions to save the higher magnitude input signal at a HIGH logic level and the lower magnitude input signal at a LOW logic level. Delay circuit 409 is connected to receive the inverse of the signal SMPLB generated by the control logic. This inverted signal is designated AZ in Figure 4. The output of delay circuit 409, signal AZ1 is provided as the input to delay circuit 420 and the output of circuit 420, signal AZ2, is provided as the input of delay circuit 440. Output signal AZ3 is provided through inverter 227 shown in Figure 2 to control logic 203 and DAC 205. The inverted AZ3 signal is designated as signal CONV in the drawings. Auto-zeroing of the comparator to eliminate charge injection error which could occur during the ADC sample period is accomplished as follows. Upon conclusion of the sample period the SMPLB signal is set HIGH opening switches S320 and S321 (shown in Figure 3) and temporarily "floating" the B bus, to which the bottom plate of the capacitor array is connected. Transistor switches T411, T412, T431, T432, T451 and T452 are all in their closed position at this time. Any charge injection is now absorbed by the comparator first stage coupling capacitors C401 through C406. When the SMPLB signal is set HIGH, the input to delay circuit 409, signal AZ, goes LOW. About sixty nanoseconds later signal AZ1 goes low opening switches T411 and T412 and removing the first stage from auto-zero mode. Any differential offsets introduced by the switching of transistors T411 and T412 are amplified by amplifier 410 and absorbed by interstage coupling capacitors C428 and C429. After the passage of another sixty nanoseconds signal AZ2 goes low opening switches T431 and T432 and removing the second stage from auto-zero mode. The above sequence is then repeated for stage three. Any offset introduced by the final stage is insignificant since its effect is divided by the gain of the first two stages. It will be appreciated that with this arrangement, the comparator 207 is auto-zeroed to a value equal to one-half of the sum of the comparator supply voltage. When signal AZ3 is set LOW, signal CONV goes HIGH initiating the successive approximation search. A schematic diagram of differential amplifier 410 is provided as Figure 5. The amplifier consists of NMOS transistors T511, T512 and T515, cascode devices T513 and T514, load devices T501 and T502; and current source devices T503 and T504. Cascode devices T513 and T514 are NMOS transistors, load devices T501 and T502 are diode-connected PMOS transistors, and current source devices T503 and T504 are PMOS transistors. A resistor R521 is provided to permit the amplifier voltage to be adjusted to VDD/2 by varying the bias current generated by transistor T515. The presence of resistor R521 does not effect the dynamic performance of the amplifier since the current flow through the resistor is always constant. Amplifiers 430 and 450 are identical in construction and operation to amplifier 410 shown in Figure 5. Figures 6A through 6C are a schematic diagram of the comparator bias circuit 470 shown in Figure 4. Referring to Figure 6A, the circuit includes transistors T609 through T613, T628 and T629 configured to form a simple operational amplifier. The negative, or inverting, input to this opamp is connected to receive signal VR50 from a tap to the midpoint of a PMOS voltage divider consisting of four PMOS transistors, labeled T605 through T608, connected in series between voltage sources VDD and VSS. The positive, or non-inverting input to the opamp is connected to output OUTM4 of a differential amplifier 661, similar to the amplifier shown in Figure 5, hard-wired in auto-zero mode, i.e. with outputs OUTM4 and OUTP4 connected to inputs INP4 and INM4, respectively. The output of the opamp, signal NBIAS, is provided through a capacitor C662 to input INM4. The opamp and differential amplifier 661 operate to establish a value for NBIAS necessary to force amplifier output OUTM4 to have a voltage of VDD/2. Circuitry for generating signals PBIAS, PB, VCASC and VR50 is shown in Figures 6B and 6C. Bias signal PBI is generated by a diode-connected PMOS transistor T601 having its source connected to supply voltage VDD and its drain connected to the drain of an NMOS transistor T621. The source of transistor T621 is connected to supply voltage VSS and its gate is connected to receive bias signal NBIAS. Signal PB, which is used to bias the delay cells (shown as elements 409, 420 and 440 in Figure 4), is provided at the gate/drain of transistor T601. PBIAS is generated by a resistor R663, diode connected PMOS transistor T604 and NMOS transistor T623 connected in series between supply voltages VDD and VSS. Resistor R663 is connected between supply voltage VDD and the source of transistor T604. Transistor T623 has its drain connected to the drain of transistor T604, its source connected to supply voltage VSS and its gate connected to receive signal NBIAS. Signal PBIAS is provided at the gate/drain of transistor T604. VCASC is generated by circuitry including a PMOS transistor T602 having its source connected to supply voltage VDD and its gate connected to receive signal PB; a diode-connected NMOS transistor T626 having its drain connected to the drain of transistor T602; a diode-connected transistor T625 having its drain connected to the source of transistor T626; an NMOS transistor T622 having its drain connected to the source of transistor T625, its source connected to supply voltage VSS and its gate connected to receive signal NBIAS; an NMOS transistor T627 having its drain connected to supply voltage VDD and its gate connected to the drain of transistor T602; and an NMOS transistor T624 having its drain connected to the source of transistor T627, its gate connected to receive signal VR50, and its source connected to the drain of transistor T622. A pair of resistors R664 and R665 are connected in series between the source and drain of transistor T626. Signal VCASC is provided at the node between resistors R664 and R665. The current in the differential transistor pair T624 and T625 is forced in balance by transistor T602, which generates a current equal to one-half of the current in transistor T622. This causes the voltage at the source of transistor T626 to be equal to VR50 and VCASC to be equal to VR50 plus on-half the voltage across transistor T626. This voltage level results in approximately equal drain-to-source voltages on transistors T621/T623 and T622/T624. Transistor T627 is used to force the drain voltages of transistor T624 and T625 to be approximately equal. Transistors T641, T642, T651, T652 and T653 are responsive to power down signals PD and PDB (which is the inverse of signal PD) to shut down operation of the bias circuit. The bias circuit, described above, auto zeros the DC voltage of each comparator stage to VDD/2. During the ADC sample period the comparator stage one input capacitors are charged to VR1 - VDD/2. At the initiation of the successive approximation sequence the voltage at the plus input to the first stage differential amplifier (element 410 of Figure 4A) will be VR1 - VIN + VDD/2 and the voltage at the minus input will be VR1 - VDD/2, where VR1 = (VREFP + VREFM)/2 and VIN can be any voltage between and including VREFM and VREFP. Therefore, reference voltages VREFP and VREFM can be set to define any voltage range within or equivalent to the range defined by comparator supply voltages VDD and VSS and all of the comparator internal nodes will remain within the supply voltage rails. Figure 7 is a schematic diagram of delay circuit 409 shown in Figure 4A. The delay circuit includes PMOS transistors T701 and T702 and NMOS transistors T711 and T712 connected in series between supply voltages VDD and VSS. Transistor T701 receives bias signal PB at its gate, transistor T712 receives bias signal NBIAS at its gate, and transistors T702 and T711 receive auto-zero signal AZ at their gates. A 0.57 pF capacitor C750 is connected between the node between transistors T702 and T711 and supply voltage VSS. Also connected to this node is the first of three series-connected inverters consisting of transistor pairs T703 and T713, T704 and T714, and T705 and T715, respectively. The output of the delay circuit is provided by the last inverter. A nominal delay of about 60 nanoseconds is generated by the delay circuit. Transistors T701, T702, T711 and T712 form a switched current source and sink which which charge and discharge capacitor C750. Inverter T703/T713 provides voltage gain while inverters T704/T714 and T705/T715 sharpen the edges of the output transitions of the delay circuit. Delay circuits 420 and 440 (shown in Figures 4A and 4B) are identical in construction to delay circuit 409. Since the comparator gain stages and the delay circuits are biased from a common bias circuit, the auto-zero response time of each comparator stage will track the delay through each delay circuit with respect to supply voltage, process variations and temperature. It can thus be seen that there has been provided by the present invention an analog-to-digital converter including a novel digital-to-analog converter and comparator implementation which permits the use of rail-to-rail references and input signals. The ADC can achieve higher voltage ranges than prior at ADCs without sacrificing accuracy. From the foregoing specification it will be clear to those skilled in the art that the present invention is not limited to the specific embodiment described and illustrated and that numerous modifications and changes are possible without departing from the scope of the present invention.

Claims: A charge redistribution amalog-to-digital conversion method for converting an unknown analog voltage (VIN) having a value between a first and second supply voltage to a binary signal by means of a capacitor array (303) formed of a plurality of capacitors each having a first and second plates, the first plates being connected to comparison means (207) which is supplied by way of a supply voltage span (Vdd, Vss), comprising the steps of: (a) generating a reference voltage (VR1) which is dependent upon the values of said first and second supply voltages by dividing said first and second supply voltages such that the reference voltage (VR1) has a value equal to one half the sum of said first and second supply voltages and supplying said reference voltage to said first plates of the capacitors; (b) providing said unknown analog voltage to the second plates of the capacitors for subtracting said unknown analog voltage (VIN) from said reference voltage (VR1), (c) auto-zeroing the DC voltage of the comparison means (207) to a value equal to half the sum of the voltages (Vdd, Vss) forming the said supply span (d) simultaneously isolating said first and second plates from said reference voltage and said unknown analog voltage respectively; (e) providing a known voltage initially equal to one half the sum of said first and second supply voltages and generating a voltage equal to the sum of the output of said subtracting step and said known voltage; (f) comparing the voltage determined by the step (e) with said reference voltage (VR1); (g) generating a new value for said known voltage (VR1) dependent upon the result of the comparison; and (h) repeating steps (e) through (g) until the voltage generated in step (e) converges to said reference voltage (VR1); A charge redistribution analog-to-digital converter for converting an unknown analog voltage to a binary signal by a method according to Claim 1, including a capacitor array (303) formed of a plurality of capacitors each having a first and second plate, the first plate being connected to comparison means (207) which is supplied by way of a supply voltage span (Vdd, Vss), reference voltage generating means (305) adapted to generate a reference voltage (VR1) which is dependent upon the value of said first and second supply voltages by dividing said first and second supply voltages such that the reference voltage (VR1) is equal to one half the sum of said first and second supply voltages, and which reference voltage generating means (305) is connected to supply said reference voltage (VR1) to said first plates of said capacitor array (303), said capacitor array (303) forming add/subtract means adapted to receive said unknown analog voltage at the second plates thereof and to subtract said unknown analog voltage (VIN) from the said reference voltage (VR1) to provide a difference voltage (VR1 - VRN) auto-zeroing means arranged for auto-zeroing the DC voltage of the comparison means (207) to a value equal to half the sum of the voltages (Vdd, Vss) forming the said suppply span, and including switching means controlled to simultaneously isolate the first and second plates from the reference voltage and said unknown analog voltage respectively, and further including voltage generating means (R301-R316, S301-S327) adapted to generate a voltage equal to the sum of the output of the add/subtract means (303) and a known analog voltage initially equal to one half the sum of said first and second supply voltages, wherein said comparison means (207) is arranged to compare the output of said voltage generator means with said reference voltage (VR1), and control means are provided for controlling the repeated generation of said voltage equal to the sum of the output of the add/subtract means (303) and a known voltage until the generated voltage converges to said reference voltage (VR1). An analog-to-digital converter according to Claim 2, characterized in that said voltage generating means includes a resistor string (R301-R316) having first and second ends connected to receive said first and second supply voltages, respectively, said reference voltage generating means including a tap (305) from the mid-point of said resistor string (R301-R316). An analog-to-digital converter according to Claim 2 or 3, characterized in that said comparison means (207) includes a plurality of series-connected comparator stages including respective differential amplifiers (410, 430, 450). An analog-to-digital converter according to Claim 4, characterized in that said comparison means (207) includes auto-zero control means (409, 420, 440) adapted to successively auto-zero said series-connected comparator stages.

Inventors: BITTING RICKY F, BITTING, RICKY F.
Applicants: HYUNDAI ELECTRONICS AMERICA, NCR INT INC, SYMBIOS INC, NCR INTERNATIONAL, INC., SYMBIOS, INC.
Application year: 1991
Main classification: H03M1
IPC codes: H03M1