Semiconductor digital circuits

Abstract: Two switches (QP23,QN23) are provided to a semiconductor circuit which includes cascade connected first circuit section and second circuit section, each having an inverter (21,22) formed by complementary transistors (QP21,QN21;QP22,QN22) connected between a power terminal (BS1,BS2) and a grounding terminal. The first switch (QN23) is inserted between the first circuit section and the second circuit section and forms a current path of the electrical charge during the state transition of the circuit. The second switch (QP23) is inserted between the input end and the power terminal of the second circuit section and controlled by the output signal of the second circuit section.

Description: BACKGROUND OF THE INVENTION Field of the Invention The present invention relates to semiconductor digital circuits, and more particularly to level conversion circuits added to semiconductor circuits. Especially, the present invention relates to semiconductor digital circuits that are formed by connecting two inverters, each consisting of transistors that are complementarily connected between a power supply terminal and a grounding terminal, which are operated at different supply voltages. Description of the Prior Art In recent years, the short channeling has increasingly been advanced in semiconductor integrated circuits such as a dynamic RAM (DRAM), and by now the development has reached even to cover the submicron region. Accompanying this trend, such problem as the deterioration in the characteristics of the elements due to the implantation of the hot carriers into the transistor has become of concern, and the reduction of the supply voltage to the semiconductor integrated circuits has been implemented as a solution to this problem. However, for connecting such a semiconductor integrated circuit to an existing peripheral circuit it becomes necessary to provide an interface mechanism which converts a low voltage to a high voltage. A conventional level conversion circuit with the aforementioned kind of interface mechanism comprises a first and a second inverters each consisting of transistors that are complementarily connected between a power terminal and a grounding terminal, that are operated at different voltages. The output of a semiconductor digital circuit with a low logic swing level is applied to the input of the first inverter for low supply voltage, and the output of the first inverter is applied to the input of the second inverter for high supply voltage. An output signal with high logic swing level is taken out from the output of the second inverter. For the second inverter it is normal to directly apply an external supply voltage through a terminal of the external power supply, and for the first inverter it is normal to apply the externally supplied source voltage after lowering it with an internal voltage reduction circuit. In such a prior art semiconductor digital circuit, it is necessary to apply to the input of the second inverter a signal with a supply voltage which is higher than the voltage value obtained by subtracting the absolute value of the threshold voltage of the transistor that is connected to the supply voltage from the external supply voltage, in order to let the output of the second inverter to make a correct transition from the external supply voltage to zero volt. The reason for this is that if the above-mentioned condition is not fulfilled, the complementarily connected two transistors that constitute the second inverter are both turned on, with the low level of the output signal not going to zero volt, a through current steadily flows continuously in the second inverter, so that the power consumption will be increased. In addition, there was a problem that the setting range of the supply voltage for the first inverter is limited. JP-A-63 39294 and US-A-4 926 070 published May 15, 1990 disclose a voltage level converting circuit with a first and a second inverting section, a first switch connected between an output end of the first invertor section and an input end of the second inverting section, and a second switch connected between the input of the second inverting section and a power terminal. The second switch is controlled by an output signal of the second inverting section. In the voltage level converting circuit of these documents, through currents may flow, at first from the power terminal through the second switch and the first inverting section to the ground line when the input of the second inverting section is discharged, and secondly from the power terminal through the second inverting section to the ground line when the output signal rises. DE-OS-33 40 567 discloses a level converting circuit according to the precharacterizing portion of claim 1. Also in this document, the through current may flow from the transistor connected to the input note to the second section to a ground line of the first circuit section. This level converting circuit, therefore, causes a large through current, making the current consumption large. Additionally, this circuit takes a long time for discharging the input node for the second circuit section, decreases the operation speed for inverting its output signal, and thus, does not achieve a high speed operation with low current consumption. US-A-4 656 373 discloses a further level converting circuit having nearly the same construction as in DE-OS-33 40 567, mentioned above. Therefore, the same problems may arise in this circuit. JP-A-62 246925 discloses a delay circuit to delay an output signal and supply a delayed signal to charging and discharging transistors, each selectively charging and discharging the output terminal so as to rise the operating speed of the circuit. The charging transistor charges the output node in addition to an inverter when the output signal is to rise. The discharging transistor discharges the node when the output signal is too small. Thus, the operating speed of the circuit is improved. BRIEF SUMMARY OF THE INVENTION Object of the Invention The object of the present invention is to further improve the transmition characteristics, and to prevent unnecessary high through currents. This object is achieved by a semiconductor digital circuit according to claim 1. The dependent claims are related to different advantageous aspects of the present invention. It is an advantage of the present invention to provide semiconductor digital circuits which can prevent the generation of a steady through current and have low power consumption. It is preferable that the semiconductor digital circuit in accordance with the present invention comprises a first circuit part having an inverter formed by connecting complementary transistors between a first power terminal and a grounding terminal, a second circuit part having an inverter formed by complementarily connecting transistors between a second power terminal and a grounding terminal and is operated at a supply voltage higher than the supply voltage of the inverter of the first circuit part, and three switches. The first switch is inserted between the first circuit part and the second circuit part, and forms a charging/ discharging path of the electric charge during the transition of the state. The second switch is inserted between the input end and the second power terminal of the second circuit part, and its conduction is controlled by the output signal of the second circuit part. The third switch is inserted between the output end and the grounding terminal of the second circuit part, and its conduction is controlled by the output signal of the first circuit part so as to output it by inverting. A fourth switch may be inserted in parallel with the third switch between the output end and the grounding terminal of the second circuit part, and the fourth switch may be controlled by the delayed signal of the output end of the second circuit part. Further, it is preferable that the semiconductor digital circuit in accordance with the present invention has a switch which is serially connected to the second switch so as to have its output end connected to the input end of the second circuit part, and its shut-off is controlled by the input signal to the first circuit part. Moreover, it is preferable that in the semiconductor digital circuit in accordance with the present invention the complementarily connected transistors are a P-channel MOS transistor and an N-channel MOS transistor. Brief Description of the Drawings The above-mentioned and other objects, features and advantages of this invention will become more apparent by reference to the following detailed description of the invention taken in conjunction with the accompanying drawing, wherein: FIG. 1 shows an example of the prior art semiconductor digital circuit; FIG. 2 shows an example of another prior art the semiconductor digital circuit; FIG. 3 shows an example of another prior art the semiconductor digital circuit; FIG. 4 shows a first embodiment of the semiconductor digital circuit of the present invention; FIG. 5 shows a second embodiment of the semiconductor digital circuit of the present invention; FIG. 6 is a graph for displaying the effect of the semiconductor digital circuit of the present invention; FIG. 7 shows a non-claimed enhancement of the circuit shown in figure 2 ; and FIG. 8 shows a further non-claimed enhancement of the circuit shown in figure 2. DETAILED DESCRIPTION OF THE INVENTION Prior to proceeding to the description of the embodiments of the present invention, a description of the prior art semiconductor digital circuits are in order. FIG. 1 shows an example of the prior art semiconductor digital circuit. This circuit includes a CMOS inverter 11 which is a first circuit part and a CMOS inverter 12 which is a second circuit part where these inverters are connected in cascade fashion. The CMOS inverter 11 is constituted of a P-channel MOS transistor QP₁₁ and an N-channel MOS transistor QN₁₁ that are complementarily connected in pair between a first power terminal BS₁ and a grounding terminal, and it inverts an input signal IN and outputs its logical output to a node N₁₁ which is its output end. Further, the CMOS inverter 12 is constituted of a P-channel MOS transistor QP₁₂ and an N-channel MOS transistor QN₁₂ that are complementarily connected in pair between a second power terminal BS₂ and a grounding terminal, and it outputs an output signal OUT by inverting the logical output that was output to the node N₁₁. Here, if V C1 is a first supply voltage applied to the first power terminal BS₁ and V C2 is a second supply voltage applied to the second power terminal, then the second supply voltage V C2 is a supply voltage applied externally to the semiconductor digital circuit and the first supply voltage V C1 is a voltage obtained by lowering the second supply voltage V C2 inside the semiconductor digital circuit. In this level conversion circuit, when the input signal IN changes from a high level (V C1 ) to a low level (0V), the voltage at the node N₁₁ changes from 0V to V C1 . In response to this, the output of the CMOS inverter 12 changes from a high level (V C2 ) to a low level (0V). In the prior art level conversion circuit described above, in order for the output of the CMOS inverter 12 to change from the V C2 level correctly to 0V, it is necessary that the following Eq. (1) is satisfied. V C1 > V C2 - |V TP |, where V TP is the threshold voltage of the transistor QP₁₂. If the above condition fails to be satisfied, both of the transistors QP₁₂ and QN₁₂ are energized, the low level of the output OUT does not go to 0V and a through current continues to flow steadily, thereby increasing the power consumption of the circuit. Because of this, there also was a problem that the setting range of the first supply voltage V C1 has to be limited. FIG. 2 shows a level conversion circuit in accordance with another prior art. This circuit includes a CMOS inverter 21 as a first logic part which is operated by the first supply voltage V C1 that is applied to the first power terminal BS₁ and a CMOS inverter 22 as a second logic part which is operated by the second supply voltage V C2 that is applied to the second power terminal BS₂, and these inverters are cascade connected. The CMOS inverter 21 consists of a P-channel MOS transistor QP₂₁ and an N-channel MOS transistor QN₂₁ whose gates are commonly connected to the input end and their drains are commonly connected to the output end, with the source of the transistor QP₂₁being connected to the first power terminal BS₁. and the source of the transistor QN₂₁ being grounded. Further, the CMOS inverter 22 consists of a P-channel MOS transistor QP₂₂ and an N-channel MOS transistor QN₂₂, with the source of the transistor QP₂₂ being connected to the second power terminal BS₂ and the source of the transistor QN₂₂ being grounded. An N-channel MOS transistor QN₂₃ is inserted between a node N₂₁ which is the output end of the CMOS inverter 21 and a node N₂₂ which is the input end of the CMOS inverter 22. The gate of the transistor QN₂₃ is connected to the first power terminal BS₁. Further, a P-channel MOS transistor QP₂₃ as a second switch is inserted between the second power terminal BS₂ and the node N₂₂. The gate of the transistor QP₂₃ is connected to the output end of the CMOS inverter 22. In the above circuit, when an input signal IN changes from a high level (V C1 ) to a low level (0V), the signal at the node N₂₁ changes from a low level (0V) to a high level (V C1 ). At this time, the source side of the transistor QN₂₃, namely, the node N₂₂, changes from a low level (0V) to a high level (V C1 - V TN ), where V TN represents the threshold voltage of the transistor QN₂₃. In order to cause the output state of the CMOS inverter 22 change from the high level to the low level in this case, it is necessary for the potential V C1 - V TN of the node N₂₂ to exceed the logic threshold voltage V TL of the CMOS inverter 22. That is, it is necessary to adjust in the design stage the gain constants of the transistors QP₂₂ and QN₂₂ so as to satisfy the relation given in Eq. (2) below. V TL < V C1 - V TN More specifically, the logic threshold voltage V TL need be set at a low value by setting the gain constant of the transistor QN₂₂ to be larger than the gain constant of the transistor QP₂₂. By so doing it is possible to realize the extension of the operating margin. When the potential of the node N₂₂ exceeds V TL , the output signal OUT changes from the V C2 level to the low level. In this process, when the level of the output signal OUT is reduced from V C2 by the amount corresponding to the threshold voltage of the transistor QP₂₃, the transistor QP₂₃ starts to be energized so that the potential of the node N₂₂ is raised further until eventually it is elevated to the level of V C2 . As a result, the transistor QP₂₂ which constitutes the CMOS inverter 22 goes to a completely de-energized state, preventing the steady flow of a through current. It should be noted that when the potential of the node N₂₂ is raised to the level of V C1 , the transistor QN₂₃ goes to a de-energized state so that the eventual potential V C2 of the node N₂₂ will never be applied directly to the CMOS inverter 21. On the other hand, when the input signal IN changes from a low level (0V) to a high level (V C1 ), the potential of the node N₂₁ changes from a high level (V C1 ) to a low level (0V). Since the transistor QN₂₃ is energized as a result of this, the potential of the source side, namely, the node N₂₂, changes from a high level (V C2 ) to a low level (0V), and the level of the output signal OUT of the CMOS inverter 22 changes from a low level (0V) to a high level (V C2 ). In response to this, the transistor QP₂₃ is de-energized. FIG. 3 is a circuit diagram of a level conversion circuit in accordance another prior art. The difference of this circuit from the circuit shown in FIG. 2 resides in that an N-channel MOS transistor QN₃₄ is connected anew to the output end of the circuit. The drain of the transistor QN₃₄ is connected to the output end of a CMOS inverter 32, its source is grounded and its gate is connected to the output end a CMOS inverter 31. The remaining constitution of the circuit is the same as that in FIG. 2, with the CMOS inverters 31 and 32, the transistors QP₃₁, QP₃₂, QP₃₃, QN₃₁, QN₃₂ and QN₃₃ and the nodes N₃₁ and N₃₂ correspond to the CMOS inverters 21 and 22, the transistors QP₂₁, QP₂₂, QP₂₃, QN₂₁, QN₂₂ and QN₂₃ and the nodes N₂₁ and N₂₂, respectively. Accordingly, the description of these parts of the present embodiment will be omitted. In accordance with the circuit of Fig. 3, the deviation between the rising characteristic and the falling characteristic of the input signal IN is small compared with that of the first embodiment so that it is possible to improve the transmission characteristic of the circuit. Namely, in the circuit of Fig. 2, the characteristic at the time of falling of the output signal OUT is arranged to be improved by lowering the threshold voltage V TL of the CMOS inverter 22. However, in accordance with the circuit of Fig. 3, the falling characteristic of the output signal OUT is improved by directly driving the transistor QN₃₄ with the output of the CMOS inverter 31, so that it is possible to realize an improvement of the transmission characteristic by raising the logic threshold of the CMOS inverter 32 to the conventional value. Next, referring to Fig. 4 to Fig. 8, the embodiments of the present invention will be described. FIG. 4 is a circuit diagram of the level conversion circuit in accordance with the first embodiment of the present invention. The difference of the circuit of the present embodiment from the circuit of Fig. 3 resides in that a P-channel MOS transistor QP₄₄ is inserted anew in addition to a P-channel MOS transistor QP₄₃ between the input end and the second power terminal 352 of a post-stage CMOS inverter 42. The drain of the transistor QP₄₄ is connected to the input end of the CMOS inverter 42, its source is connected to the drain of the transistor QP₄₃ and its gates is connected to the input end of a CMOS inverter 41. The remaining constitution is the same as that in FIG. 3, and the CMOS inverters 41 and 42, the transistors QP₄₁, QP₄₂, QP₄₃, QN₄₁, QN₄₂, QN₄₃ and QN₄₄, and the nodes N₄₁ and N₄₂ correspond to the CMOS inverters 31 and 32, the transistors QP₃₁, QP₃₂, QP₃₃, QN₃₁, QN₃₂, QN₃₃ and QN₃₄, and the nodes N₃₁ and N₃₂, respectively. Accordingly, the description on these parts of the present embodiment will be omitted. In accordance with the present embodiment it is possible to further improve the transmission characteristic at the time of rising of the output signal OUT than that of the second embodiment. Namely, in the circuit of Fig. 3, when the input signal IN goes to a high level, the output signal OUT is at a low level, so that there is a current which flows out of the second power terminal BS₂ and flows into the transistor QN₃₁ via the transistors QP₃₃ and QN₃₃, and it takes some time for the discharging at the nodes N₃₁ and N₃₂. In contrast, according to the circuit of the present embodiment, there is inserted a transistor QP₄₄ in the above-mentioned current path which goes to an interrupted state at the time of rise of the input signal IN so that it is possible to improve the transmission characteristic. FIG. 5 is a circuit diagram of the level conversion circuit in accordance with the second embodiment of the present invention. The difference of the circuit of this embodiment from the circuit of the first embodiment resides in that an N-channel MOS transistor QN₅₅ is inserted anew between a transistor QN₅₄ and a grounding of the output stage, and the output signal OUT is input to the gate of the transistor QN₅₅ via a delay circuit 53. The remaining constitution is the same as that in FIG. 4, and the CMOS inverters 51 and 52, the transistors QP₅₁, QP₅₂, QP₅₃, QP₅₄, QN₅₁, QN₅₂, QN₅₃ and QN₅₄, and the nodes N₅₁ and N₅₂, correspond to the CMOS inverters 41 and 42, the transistors QP₄₁, QP₄₂, QP₄₃, QP₄₄, QN₄₁, QN₄₂, QN₄₃ and QN₄₄, and the nodes N₄₁ and N₄₂, respectively. Accordingly, the description about these parts of the present embodiment will be omitted. In the present embodiment, the transistor QN₅₅ is turned off following a lapse of a predetermined length of time after a fall of the output signal OUT, so that the series circuit of the transistors QN₅₄ and QN₅₅ is energized only at the instant of the fall of the output signal OUT, which acts to lower the logic threshold. Because of this, it is possible to improve the resistance to the noise by preventing the malfunction of the CMOS inverter 52 due to the voltage drop within the integrated circuit after the change of the output signal OUT. As described in the above, in accordance with the present invention the second switch is energized in response to the output change of the second circuit part, making it possible to use the signal at the input end of the second circuit part as the second supply voltage level, so that there will be no flow of a steady through current in the second circuit part and a reduction in the power consumption can be realized. Moreover, it is possible to extend the setting range of the first supply voltage. FIG. 6 is a graph illustrating the dependence of the dc current in the driver on the gate width taken of the prior art, presented for the purpose of showing the effect of the present invention. The conventional model used in the above has the same circuit as that shown in FIG. 1, and the various constants are set as in the following. WP₁ : WN₁ = WP₂ : WN₂ = 2:1, WP₂ = 3WP₁, L P = 1.0 µm, L N = 0.8 µm, V C1 = 3.3 V, V C2 = 5 V. In the above equations, WP₁, WP₂, WN₁ and WN₂ are the gate widths of the P-channel MOS transistor QP₁₁, the P-channel MOS transistor QP₁₂, the N-channel MOS transistor QN₁₁ and the N-channel MOS transistor QN₁₂, respectively, and L P and L N are the gate lengths of the P-channel MOS transistors QP₁₁ and QP₁₂, and the N-channel MOS transistors QN₁₁ and QN₁₂, respectively. As can be seen from the figure, in the conventional level conversion circuit there is a flow of a through current in the range of 10 µA to 100 µA depending upon the dimensions of the transistors used. In contrast, this current can be made equal to zero under the identical conditions according to the circuits of Fig. 2 and Fig. 3 and according to the present invention. It should be noted that the description has been presented in conjunction with the case of using the CMOS inverters for the first and the second circuits parts in each of the aforementioned embodiments, but it is needless to say that the present invention can also be applied to other logic gates. For example, FIG. 7 shows the case of replacing the first stage MOS inverter 21 in FIG. 2 by a NAND gate 71. The second stage of FIG. 7 remains the same as in FIG. 2. The NAND gate 71 is constituted of two P-channel MOS transistors QP₇₁ and QP₇₂ and two N-channel MOS transistors QN₇₁ and QN₇₂. This example corresponds to the case where the output stage of an internal digital circuit with small logic swing level is constituted not of an inverter but of a NAND gate, and the level of its output is connected by a second stage inverter. Moreover, in the circuit shown in FIG. 8 the first stage MOS inverter 21 in FIG. 2 is replaced by two MOS inverters (81 and 82), and the second stage MOS inverter 22 is replaced by a NAND gate 83. The NAND gate 83 consists of two P-channel MOS transistors QP₈₃ and QP₈₄, and two N-channel MOS transistors QN₈₃ and QN₈₄. This represents an example where the second stage inverter is served for both of a level converter and a NAND gate for the output stage of an internal digital circuit.

Claims: A semiconductor digital circuit comprising: a) a first circuit section (41) having complementary transistors connected between a first power terminal receiving a first voltage and a grounding terminal; b) a second circuit section (42) having complementary transistors connected between a second power terminal receiving a second voltage different from said first voltage and said grounding terminal; c) a first switch (QN43) connected between an output end of said first circuit section and an input end of said second circuit section and operatively serving as a current path of the electric charge therebetween during a transition state of the circuit; and d) a second switch (QP44) connected between said input end of said second circuit section and said second power terminal and controlled by an output signal of said second circuit section, characterized by further comprising e) a third switch (QP43) connected in series with said second switch between said input end of said second circuit section and said second power terminal and controlled by an input signal supplied to said first circuit section. A circuit according to claim 1, further comprising a fourth switch (QN44) connected between an output end of said second circuit section and said grounding terminal and controlled by the output signal of said first circuit section. A circuit according to claim 2, further comprising a fifth switch (QN55) connected in series with said fourth between said output end and said grounding terminal of said second circuit section, and a delay circuit (53) for delaying said output signal of said second circuit section to produce a delayed signal, said fifth switch being controlled by said delayed signal. A circuit according to claim 3, wherein a series connection of said fourth and said fifth switches turns to a conductive state to make a current path between said output end of said second circuit section and said grounding terminal when said output signal of said second circuit section falls to a ground voltage of said grounding terminal. A circuit according to claim 1, wherein said first voltage is lower than said second voltage.

Inventors: KOMURO TOSHIO, KOMURO, TOSHIO, KOMURO, TOSHIO, C/O NEC CORPORATION
Applicants: NIPPON ELECTRIC CO, NEC CORPORATION
Application year: 1991
Main classification: H03K19
IPC codes: H03K3, H03K19