Sampling rate converter

Abstract: The apparatus converts a sampling frequency of digital signals sampled at a first sampling frequency into a second sampling frequency. It produces output clocks (CK D2 ) corresponding to the second sampling frequency, which are counted in a cyclic fashion from an initial value to a maximum value based on periodicity of the first and second sampling frequencies. Coefficient addresses (D COE1 ) are generated in accordance to the resulting count value (D CNT1 ). A coefficient correction value (D HC ) is generated in accordance with a number of counts of the maximum value (S CO ) and the periodicity of the first and second sampling frequencies, and the number of the maximum value, which is a number of times of counting to the maximum value. The sampling frequency of digital signals sampled at the first sampling frequency is thus converted into the second sampling frequency with a simple circuit based on counting to the maximum value in accordance to the periodicity of the sampling frequencies.

Description: BACKGROUND OF THE INVENTION This present invention relates to a sampling rate converter, and is suitably applied to a case where a sampling frequency of component digital video signals based on a D-1(625/50) format is converted to a sampling frequency corresponding to composite digital video signals based on a D-2 format, for example. It has been known to use a sampling rate converter constituted by a digital filter for converting digital signals, obtained by sampling analog signals at a predetermined frequency, to an arbitrary sampling frequency. Such a sampling rate converter is constituted by a high order oversampling filter to strictly observe the Nyquist frequency as a conversion characteristic of the transmission system. FIGS. 1A-1D, approximately illustrate the operation of a sampling rate converter constructed by an oversampling filter. In such an oversampling filter, input digital signals (indicated by character o in FIG-1A) are inputted to the oversampling filter, the input digital signals being provided by sampling imaginary analog signals S VT as shown in FIG. 1A at a first sampling frequency f₁ (FIG. 1B). Subsequently, the oversampling filter oversamples input digital signals at a frequency f₁₁ 11 times larger, as shown in FIG. 1C , and resamples them at a second sampling frequency f₂ (FIG. 1D) to obtain output digital signals. The latter have values indicated by characters □ in FIG.1A and are rate converted into frequency f₂. When a sampling frequency of component digital video signals based on the D-1 (625/50), format as used in a digital video tape recorder (DVTR) is converted into a sampling frequency corresponding to PAL composite digital video signals based on the D-2(PAL) format, it is not possible to directly convert sampling frequency between digital video signals since the sampling frequency is rate converted from a frequency 13.5 MHz to a frequency 17.734475 MHz. It is thus necessary to build an oversampling filter having approximately a length of the order of about 16500. Also, in the converse case where a sampling frequency of PAL composite digital video signals is converted into a sampling frequency corresponding to 625/50 component digital video signals, it is necessary to build an oversampling filter in the same circuit scale as the above described case since the sampling frequency is rate converted from a frequency 17.734475 MHz to a frequency 13.5 MHz. It is thus inevitable that the overall circuit scale becomes complicated and large. To solve such a problem, a sampling rate converter has been proposed in which a oversampling filter of reduced circuit scale is built whereby the relationship of sampling frequencies of 13.5 MHz and 17.734475 MHz, between 625/50 component digital video signals and PAL composite digital video signals, are used to set an oversampling frequency, and to enter variable filter coefficients to one multiplier corresponding to a magnification of the oversampling to weight the sampling data. FIG. 2 illustrates a basic construction of a sampling rate converter 1 having changeable coefficients. In this example the sampling rate converter as shown in FIG. 2, converts a sampling frequency of 625/50 component digital video signals into a sampling frequency corresponding to PAL composite digital video signals, (for example, 625/50). Component digital video signals S IND1 and output clock signals CK D1 corresponding to the sampling frequency thereof are inputted to the timing adjusting circuit 2. The timing adjusting circuit 2 outputs data of the 625/50 component digital video signals S IND1 , which is inputted at the rate of the clock signal CK D1 at the timing of the rate of the clock signal CK D2 so as to exchange the clock as well as control the timing of the data transference described below. In addition to this, frame pulses P FD1 corresponding to one frame of 625/50 component digital video signals S IND1 are inputted to reset terminals of a phase locked loop (PLL) circuit 3 and a counter 4. The output of PLL circuit 3 is fed back through a frequency divider 5 with a dividing ratio (= 1/709379) corresponding to the number of samples of one frame (= 709379) of PAL composite digital video signals. This enables the PLL circuit 8 to correctly synchronize with frame pulses P FD1 of 625/50 component digital video signals, and the PLL circuit 3 generates output clock signals CK D2 corresponding to the sampling frequency of the PAL composite digital video signals and supplies them to a count input terminal of the counter 4, the timing adjusting circuit 2 and an oversampling filter 6. The counter 4 is reset according to frame pulses P FD1 , and counts output clock signals CK D2 , inputted from the PLL circuit 3, at every frame. Consequently, count data D CNT varying from zero to 709378 is sequentially sent to a coefficient address control circuit 7. The coefficient address control circuit 7 generates coefficient address data D COE according to the count data D CNT to read coefficients C OEFA , C OEFB , ... from coefficient generating circuits 9A, 9B, ... which are ROM (read only memories), the coefficients C OEFA , C OEFB , ... are fed to multipliers 8A, 8B, ... which serve as weighting means of the oversampling filter 6, of which part is illustrated in FIG. 3. The relationship between coefficients C OEFA , C OEFB , ... supplied to the multipliers 8A, 8B, ... and the coefficient address data D COE is illustrated schematically in FIG. 4. In FIG. 4, the Y-axis represents the coefficient C OEF and the X-axis represents the coefficient address data D COE . The X-axis can also be considered as a time axis. "N" is the order of the oversampling filter and is derived from the characteristic of the oversampling filter required. An impulse response characteristic of the oversampling filter is formed by N groups of the coefficient C OEF . It should be noted that the impulse response characteristic illustrated in FIG-4, is schematic rather than realistic. "M" represents the number of divisions between sampling points to the input signal S IND1 , that is a resolution capability, by which the oversampling frequency is decided. The groups of coefficient are divided by M groups so as to store them in the coefficient generating circuits 9A, 9B, ... , of which there are M/N. The addresses a₀... a n , b₀... b n are appointed with a coefficient address data D COE generated at the rate of the clock signal CK D2 in the coefficient address control circuit so that the groups of coefficient C OEF stored can be read out. The relationship between the count data D CNT and the address of the coefficient generating circuits 9A, 9B, ... appointed with the coefficient address data D COE is illustrated in FIG. 5. In the case illustrated in FIG. 5, the oversampling filter is constructed with orders N = 4554, M = 506. The coefficients which form the impulse response characteristic of the oversampling filter are stored in sequence at the addresses from a₀, b₀, ... i₀ to a₅₀₅, b₅₀₅.The lower portion of FIG.5 illustrates the coefficient addresses D COE which are generated at the timings of each of the count data D CNT . In FIG. 5 , with the count data D CNT exchanged in sequence n, n+1, n+2, ... at the rate of clock signal CK D2 , the coefficient address data D COE is generated so that the addresses a₀ - a₅₀₅, b₀ - b₅₀₅, c₀ - c₅₀₅, ... are apointed in sequence [a₀, b₀, c₀, ...], [a₃₈₆, b₃₈₆, c₃₈₆, ...], [a₂₆₆, b₂₆₆, c₂₆₆, ...], .... The appointment of the address in each of the coefficient generating circuits is performed on the basis of the relationship between the sampling period of 625/50 component digital video signal and the sampling period of the PAL composite digital video signal, so as to shift cyclicly by 386 addresses among addresses 0 - 505, as per the following equation. 506 x 13.5 17.7 = 386 The coefficients C OEFA - C OEFI generated in the coefficient generating circuit 9A - 9I are multiplied by the output data in the multiplier. When the count data D CNT takes on predetermined values, for instance D CNT becoming D CNT = n + 1 in FIG. 5, it is necessary to multiply a coefficient corresponding to D CNT = n + 1 by the same data as the flip-flop output data multiplied at the timing D CNT = n. In this case, the coefficient address control circuit 7 generates a shift control signal D SFT to stop the data transference in the flip-flop circuits 10A - 10I(this is achieved by shifting one data block) so as to be able to multiply in the multiplier using the same flip-flop output data. The shift control signal D SFT also is applied to the timing adjusting circuit. This results in stopping the exchange of the clock according to the stopping of the operation of the data transference from the flip-flop circuits 10A - 10I, whereas the exchange of the clock continues in the timing adjusting circuit 2. As described above, the timing adjustment circuit 2 adjusts the timing of inputted 625/50 component digital video signals S IND1 to send to the oversampling filter 6 - according to the output clock signals CK D1 - the output clock signals CK D2 , and the shift control data D SFT . The oversampling filter 6 performs oversampling by multiplying the thus inputted 625/50 component digital video signals S IND1 by coefficients C OEFA , C OEFB , ... according to the shift control data D SFT and the coefficient address data D COE , and provides outputs according to output clock signals CK D2 . In this manner, output digital video signals S OUTD2 are obtained by converting the sampling frequency of the 625/50 component digital video signals to a sampling frequency corresponding to the PAL composite digital video signals. It is to be noted that in the oversampling filter 6 input digital signals S IN are fed to a series circuit of flip-flops 10A, 10B, ... having a delay corresponding to output clock signals CK D2 . Delayed digital signals sent out from each of the flip-flops 10A, 10B, ... are inputted to subsequent flip-flops 10B, ... and to corresponding multipliers 8A, 8B,.... Predetermined coefficients C OEFA , C OEFB , ... described above are inputted from the coefficient generating circuits 9A, 9B, ... to respective multipliers 8A, 8B, ..., in order that the delayed digital signals can be mulitplied by respective coefficients C OEFA , C OEFB , .... Results of the multiplication are inputted to each of adders 11A, 11B, ... for providing a total, and the result of this addition is output as digital video signals S OUT . The sampling rate converter 1 with such a construction has a large scale circuit configuration in which the counter 4 counts a value 0 to 709378. Also, the coefficient address control circuit 7 inevitably has a large scale circuit configuration for processing count data D CNT from the counter 4. Consequently, there is a problem in that the sampling rate converter 1 becomes large sized. SUMMARY OF THE INVENTION In view of the foregoing, an object of this invention is to provide a sampling rate converter which is capable of converting a sampling frequency of digital signals sampled at a first sampling frequency to a second sampling frequency, using a simple construction. In accordance with a first aspect of this invention, there is provided a sampling rate converter 20 for converting a sampling frequency of digital signals sampled at a first sampling frequency to a second sampling frequency, the sampling rate converter including an oversampling filter 6 having coefficient generation means 9A, 9B for generating coefficients C OEFA , C OEFB to be provided to weighting means 8A, 8B according to coefficient addresses D COE1 . According to the present invention, the above problem is solved by : counting means 21 for counting output clocks CK D2 in a cyclic fashion; correction value generation means 22, 26 for generating a coefficient correction value D HC in accordance with a number of counts of a maximum value C HC and the periodicity of the first and second sampling frequencies, the number of counts of the maximum value being obtained by counting maximum value information S CO sent from the counter means 21 every time the counter means counts to the maximum value; and coefficient address generation means 23, 24, 27 for generating the coefficient addresses D COE1 in accordance with the counted value D CNT1 , inputted from the counter means 21, and the coefficient correction value D HC inputted from the correction value generation means 22, 26. In this construction, output clocks CK D2 corresponding to the second sampling frequency are counted in the cyclic fashion from the initial value to the maximum value based on periodicity of the first and second sampling frequencies. Coefficient addresses D COE1 are generated in accordance to the resulting count value D CNT1 and the coefficient correction value D HC in accordance with the number of counts of the maximum value C HC and the periodicity of the first and second sampling frequencies, the number of the maximum value counts being the number of times of counting to a maximum value. The sampling frequency of digital signals sampled at the first sampling frequency is thus converted to the second sampling frequency with a simple construction based on counting from the initial value to the maximum value. The nature, principle and utility of the invention will become more apparent from the following detailed description when read in conjunction with the accompanying drawings in which like parts are designated by like reference numerals or characters. In the accompanying drawings: FIG. 1 is a timing chart illustrating the oversampling operation according to prior art; FIG. 2 is a block diagram showing the conventional sampling rate converter; FIG. 3 is a block diagram showing part of the oversampling filter of FIG. 2; FIG. 4 is a graph showing the relationship between the coefficient and the coefficient address data in the prior art; FIG. 5 is a schematic diagram showing the relationship between the count data D CNT and the coefficient address data D COE in the prior art; FIG. 6 is a block diagram showing one embodiment of the sampling rate converter according to this invention; and FIG. 7 is a schematic diagram showing the shift of the coefficient address data with the correction according to the invention. DETAILED DESCRIPTION OF THE INVENTION Preferred embodiments of this invention will be described with reference to the accompanying drawings: (1) Principle of the Embodiment In a sampling rate converter of this embodiment, the sampling frequency of component digital video signals based on the D-1(625/50) format in a digital video tape recorder (DVTR) is converted to a sampling frequency corresponding to the PAL composite digital video signals based on the D-2(PAL) format. In practice, the sampling frequency of component digital video signals based on the D-1(625/50) format is defined to be 13.5 MHz., and the number of samples for a line of digital video signals is hence 864. The total number of samples of one frame amounts to 540,000. The sampling frequency of PAL composite digital video signals based on the D-2(PAL) format is 17.734475 MHz, which is four times the sub-carrier frequency f SC calculated based on the following equation, provided the horizontal frequency f h is 15.625 KHz: f SC = (284 - 1 4 )f h + 50 2 = 4.43361875 (MHz) Accordingly, the number of samples per a line of digital video signals is 1135.0064, and the total number of samples of one frame is 709379. As described, the ratio, in number, of samples for a line of the 625/50 component digital video signals over the PAL composite digital video signals is 864:1135.0064, and it is understood that the ratio does not have an appropriate integer relationship. When a sampling frequency is converted using an oversampling filter with a sufficient length, the oversampling filter may be of a coefficient switch type construction. In this case, the combination of coefficients given to the oversampling filter must correspond to the phase between the pre-conversion sampling frequency and the post-conversion sampling frequency. As already described, there is no relatively small integer ratio between the pre-conversion sampling frequency and the post-conversion sampling frequency, and it is hence necessary to count sampling points for a frame so as to generate a coefficient corresponding to the count value. The sampling cycle of the 625/50 component digital video signals is computed from the following equation: 1 13.5 (MHz) = 74.074074 (nsec) The sampling cycle of the PAL composite digital video signals is computed from the following equation: 1 17.734475 (MHz) = 56.38734724 (nsec) By detecting the time difference between the sampling point corresponding to the sampling frequency of 625/50 component digital video signals and the sampling point corresponding to the sampling frequency of PAL composite digital video signals from the top of a frame, the number of samples of the 625/50 component digital video signals becomes a value given by the equation below, at a time point of 423 samples of PAL composite digital video signals when counted from the top of the frame: 56.38734724 (nsec) x 423 74.074074 (nsec) = 321.9999464 the sampling points of both become closer in timing. Sampling points of both which correspond to the sampling cycles of 625/50 component digital video signals and PAL composite digital video signals become closer every multiple of 423 sample points at the sampling cycle of PAL composite digital video signals. In this embodiment, coefficient addresses and the data shift control are held between 423 sample points with reference to the sampling cycle of the PAL composite digital video signals. In this manner, correction of time given by the equation below is performed every 423 sample points at the sampling cycle of the PAL composite digital video signals. The corresponding equation can be expressed as : 56.38734724 (nsec) x 423 - 74.074074 (nsec) x 322 = - 0.00396796 (nsec) In phase shifting and generation of coefficients, the above-described correction is carried out by providing a counter which counts sampling within a small range between 423 sample points. This enables the sampling frequency to be converted at a high accuracy with a simple construction. (2) Sampling Rate Converter of the Embodiment In FIG. 6, in which corresponding parts of FIG. 2 are given the same reference numerals, 20 generally designates a sampling rate converter which converts a sampling frequency of 625/50 component digital video signals to a sampling frequency according to PAL composite digital video signals, for example. In this embodiment, frame pulses P FD1 which correspond to every one frame of 625/50 component digital video signals S IND1 are inputted to a PLL circuit 3, a reset input terminal of a counter 21, and a reset terminal of a number-of-corrections counter 22. The counter 21 is adopted in place of the conventional counter 4 . It counts to 709378 and is designed to count output clock signals CK D2 in a cyclic manner from zero to 422 on the basis of the principle stated in connection with the equations (3) to (6) during one frame of 625/50 component digital video signals S IND1 . The counter 21 thus counts output clock signals CK D2 during one frame. The counter 21 sends resulting count data D CNT , which goes from zero to 422 to a coefficient address generation circuit 23 . It also sends a carry signal S CO , which is generated every time the counted value exceeds 422, to the correction frequency counter 22. The coefficient address generation circuit 23 generates an coefficient address ADR COE and shift data D SFT1 (this signal has the same function as the shift control signal D SFT , i.e. a data transfer function and a clock exchange stopping function) in response to the inputted count data D CNT1 , and sends them to an adder 24 and a shift control circuit 25, respectively. During one frame, the number-of-corrections counter 22 counts inputted carry signals S CO to generate number-of-corrections data C HC , which is sent to a corrected value multiplier 26. The corrected value multiplier 26 calculates number-of-corrections data C HC by multiplying the correction value (= 0.00396795 (nsec)) described in connection with the equation (6) with the number-of-corrections data C HC , the corrected value data D HC corresponding to correction of the coefficient address ADR COE based on the number of corrections. The corrected value data D HC is fed to the adder 24. The adder 24 adds the coefficient address ADR COE and the corrected value data D HC to produce a corrected coefficient address ADR COE1 , which is provided to a coefficient adjusting circuit 27. On the basis of the corrected coefficient address ADR COE1 , the coefficient adjusting circuit 27 generates coefficient address data D COE1 and sends it to coefficient generating circuits 9A, 9B, ... of the oversampling filter 6. Moreover, the coefficient adjusting circuit 27 detects whether or not the coefficient address ADR COE1 is in overflow by comparison with the coefficient address ADR COE , before correction, so as to send the resultant to the shift control circuit 25 as a overflow data D OF . This overflow can be understood as follows. In FIG. 7, when the count data D CNT is D CNT = n the address a₁₃₇ is appointed as the address of the coefficient generating circuit. Subsequently, when the count data D CNT becomes D CNT = n + 1 the address a₀ is appointed by means of correction. In contrast, when the count data D CNT becomes D CNT = n + 1, the address a₅₀₅ is appointed, unless the address moves by means of correction. The overflow occurs when the address exceeds a round address where the accesses of a round address is from a₀ to a₅₀₅. If such an overflow occurs, it is necessary to process as follows. Let it be assumed that there is an address relationship as in FIG.7.The output data of the flip-flop circuit 10A to be multiplied by the coefficient having the address a₀ needs to transfer data afresh by one clock, unless that data is to stop the transfer of the data to be multiplied by the coefficient. The operation "stop of data transference" and "stop of clock exchange" described above effectively cause such a stop. The shift control circuit 25 detects whether or not there is need for "stop of data transference" and "stop of clock exchange". It sends the resultant signals as shift control signals S SFT2 to the timing adjusting circuit 2 and the oversampling filter 6. Consequently, the timing adjusting circuit 2 controls the timing of the sending of 625/50 component digital video signals S IND1 to the oversampling filter 6 according to the output clock signals CK D1 , the output clock signals CK D2 , and the shift control data D SFT2 . The oversampling filter 6 performs oversampling by multiplying the 625/50 component digital video signals S IND1 thus inputted by coefficients C OEFA , C OEFB , ... according to the shift control data D SFT2 and the coefficient address data D COE1 , and conducts resampling according to the output clock signals CK D2 and outputs the resultant data. In this manner, output digital video signals S OUTD2 which are produced by converting the sampling frequency of 625/50 component digital video signals to the sampling frequency of PAL composite digital video signals are obtained. The oversampling filter 6 has coefficients equal in number to 505 times the oversampling frequency, and hence has a resolution of 1/505. In fact, the correction (= 0.00396795 (nsec)) stated about the equation (6) is 0.027068 times as large as the resolution. This value means that correction is carried out 45 times (that is, 0.178 (nsec)) at the trailing and of one frame. To do this, the counter 21 and the number-of-corrections counter 22 are controlled to be placed in initialized states by resetting them every frame. According to the foregoing construction, sampling points are sequentially counted in a small range of 423 sampling points, and the predetermined correction is carried out every 423 sampling points. The sampling rate converter 20 is realized so that it enables the counter circuit to be built using a simple circuit construction which counts from zero to 422, and which is capable of converting a sampling frequency of 625/50 component digital video signals to a sampling frequency according to PAL composite digital video signals, at a high accuracy. Moreover, according to the foregoing construction, the coefficient address generation circuit 23 is also built with a simple circuit construction by constructing the counter 21 with a simple circuit construction which counts zero to 422. A sampling rate converter is thus realized which is capable of reducing in circuit scale to about 1/1677 of the conventional sampling rate converter 1 (FIG. 2), resulting in a remarkable reduction in overall scale. (3) Other Embodiments (3-1) In the embodiment above, a case where a sampling frequency of 625/50 component digital video signals is converted to a sampling frequency according to PAL composite digital video signals is stated but this invention is not limited to this. The invention can be suitably applied to a case where a sampling frequency of PAL composite digital video signals is converted to a sampling frequency according to 625/50 component digital video signals. It is to be understood that, in this case, an effect similar to that of the preceding embodiment will be achieved by adding the correction (= 0.00396795 (nsec)) described about the equation (6) every 322 sampling points of a sampling cycle of 625/50 component digital video signals according to the principle of the invention with reference to the equations (3) to (5). (3-2). The preceding embodiments concern cases where a sampling frequency of PAL composite digital video signals or 625/50 component digital video signals is converted to a sampling frequency according to 625/50 component digital video signals or PAL composite digital video signals. However, the invention may also be suitably applied to various cases where a sampling frequency of digital signals is converted to a new sampling frequency which does not have a simple relationship integer ratio with the sampling frequency. (3-3) In the preceding embodiments, a case where a range among every 423 sampling points of a sampling cycle of PAL composite digital video signal is counted is stated, but the invention is not limited to this. The invention can be suitably applied to a case where a range among every sampling points being integer times as many as 423 is counted.

Claims: A sampling rate converter for converting a sampling frequency of digital signals sampled at a first sampling frequency to a second sampling frequency, including an oversampling filter having coefficient generation means for generating coefficients to be provided to weighting means according to coefficient addresses, characterized in that it comprises : counting means for counting output clocks in a cyclic fashion from an initial value to a maximum value based on the periodicity of the first and second sampling frequencies, the output clocks corresponding to the second sampling frequency; correction value generation means for generating a coefficient correction value in accordance with a number of counts of the maximum value and the periodicity of the first and second sampling frequencies, the number of the maximum value counts being obtained by counting maximum value information sent from the counter means every time the counter means counts to the maximum value; and coefficient address generation means for generating the coefficient addresses in accordance with the counted value, inputted from the counter means, and the coefficient correction value inputted from the correction value generation means. A sampling rate converter according to claim 1, wherein said correction value generation means comprises: means for detecting the number of correction times, for detecting the number of times a correction occurs during one frame period, based on said maximum value information obtained from an output of said counting means; multiplication means for multiplying an output of said correction detection means by a correction value to generate said coefficient correction value. A sampling rate converter according to claim 1, wherein said coefficient address generation means comprises: coefficient address generating means for generating a coefficient address and shift data based on said count value obtained from an output of said counting means; addition means for adding said coefficient address and said correction value; coefficient adjusting means for generating said coefficient adress data based on output data of said addition means, and for detecting said coefficient address data; delayed by one cycle; which is outputted as overflow data; shift control means for generating said shift control signal based on said shift data and said overflow data. A sampling rate converter according to claim 2, wherein said correction detection means is reset with counting means by a signal representing a frame period in connection with said first sampling frequency.

Inventors: FUJITA TADAO, FUJITA, TADAO
Applicants: SONY CORP, SONY CORPORATION
Application year: 1991
Main classification: H04N7
IPC codes: H04N7, H04N9, H03H17, H04N5