A deflection driver in a video apparatus

Abstract: A driver stage (100) for a horizontal deflection output transistor (Q2) includes a transformer (T1) having a first winding (W1) coupled to an input signal (I c1 ) at a horizontal deflection frequency, a second winding (W2) coupled across a base-emitter junction of the output transistor (Q2) and a third winding (W3) coupled to the emitter of and in series with the output transistor (Q2). The third winding (W3) couples the emitter current (I c2 ) to the base of the output transistor (Q2) for supplying a substantial portion of the base current (I b2 ) during the second half of trace.

Description: The invention relates to a television deflection apparatus. US-A-4,323,825 discloses an electron beam deflection circuit useful in cathode ray tube displays which employ television scanning principles. The deflection circuit has transformer coupling between a source of horizontal drive signal and a switching transistor for the horizontal deflection coil. Energy stored in the transformer during the scan and retrace time is utilized during the scanning time to provide base current to the switching transistor. In addition, the transformer has a second primary winding connected in the emitter circuit of the switching transistor in a manner to allow positive feedback. The positive feedback enhances and sustains the base current during scanning time and controls its magnitude. Another prior art horizontal deflection circuit that has been used in various color television receivers includes a driver stage having a transformer coupled to a control terminal of the output transistor of the deflection circuit output stage. The driver stage further includes a second switching transistor having a collector electrode that is coupled to a primary winding of the transformer. When the switching transistor of the driver stage is conductive, magnetic energy is stored in the inductance of the primary winding. At the same time, a transformer coupled voltage developed across a secondary winding of the transformer causes the switching output transistor of the output stage, driven by the voltage developed in the secondary winding of the transformer, to be non-conductive. When the switching transistor of the driver stage becomes non-conductive, the stored magnetic energy in the coupling transformer produces, in a flyback manner, a winding current that flows in the secondary winding of the coupling transformer to form a forward base current for the output transistor. The winding current turns on the switching output transistor and maintains it turned-on until substantially the end of horizontal trace. The control electrode of the second switching transistor is coupled to a horizontal oscillator. When the oscillator is turned off when the television receiver enters stand-by mode, the deflector circuit output stage may self oscillate during the transition to stand-by mode. In accordance with the present invention, there is provided a television deflection apparatus, comprising: a deflection retrace resonant circuit including a deflection winding and a retrace capacitance; a first switching transistor responsive to a control signal at a frequency that is related to a deflection frequency and coupled to said retrace resonant circuit for performing a switching operation that generates a deflection current in said deflection winding; a transformer having a secondary winding that is coupled to a control terminal of said first switching transistor for developing said control signal at said control terminal; a source of an input signal at a frequency that is related to a deflection frequency; and a second switching transistor responsive to said input signal and coupled to a primary winding of said transformer for generating a first current in said second switching transistor and in said primary winding of said transformer that is transformer coupled via said transformer to said control terminal of said first switching transistor to develop said control signal at said control terminal during normal operation, characterized by means responsive to an on/off control signal which turns off said source of said input signal to initiate a standby mode of operation in which said second transistor is turned off for maintaining said second switching transistor conductive during a transition interval that precedes a change from normal operation to said standby mode of operation such that said first current flows in said primary winding and in said second switching transistor, during said transition interval. In the Drawing: FIGURE 1 illustrates a driver stage, embodying an aspect of the invention, of a horizontal deflection circuit output stage; FIGURES 2a-2f illustrate waveforms useful for explaining the operation of the circuit of FIGURE 1; FIGURES 3a-3f illustrate additional waveforms useful for explaining the operation of the circuit of FIGURE 1; FIGURE 4 illustrates a driver stage similar to that in FIGURE 1 that is used for driving a horizontal output stage of a CRT having a 90° deflection angle; FIGURE 5 illustrates a driver stage similar to that in FIGURE 1 that is used for driving a horizontal output stage having a diode modulator; FIGURE 6 illustrates a driver stage similar to that in FIGURE 1 that is used for driving a horizontal output stage having a switched E-W raster correction circuit; FIGURES 7a-7d illustrate waveforms useful for explaining the operation of each of the circuits of FIGURES 4 and 6; and FIGURE 8 illustrates a driver stage of a deflection circuit operating at 32KHZ. FIGURE 1 illustrates a driver stage 100, embodying an aspect of the invention, for driving a switching output transistor Q2 of a horizontal deflection circuit output stage 101 of a television receiver. A conventional horizontal oscillator 102, that produces, during normal power-on operation, an output signal V OSC , at a horizontal frequency f H , that is approximately 16 KHZ during normal power-on operation, is coupled via a capacitor C2 and a resistor R2 to a base electrode of a driver transistor Q1. A diode D1 coupled to the base of transistor Q1 prevents the base voltage from becoming excessively negative. A collector of transistor Q1 is coupled to one end terminal of a primary winding W1 of a coupling transformer T1. The other end terminal of winding W1 is AC bypassed to ground by a filter capacitor C1, and coupled via a current limiting resistor R1 to a source of supply voltage V+ such that resistor R1 and capacitor C1 form a ripple filter. A secondary winding W2 of transformer T1 is coupled across a base-emitter junction of transistor Q2 and across a resistor R4. A junction terminal 101a that is coupled at a junction of winding W2, resistor R4 and the emitter of transistor Q2, is coupled to one end terminal of a winding W3 of transformer T1. The other end terminal of winding W3 is coupled to ground. The collector of transistor Q2 is coupled to a damper diode D2 of a conventional horizontal deflection circuit 103. Circuitry 103 further includes a horizontal deflection winding L H , a retrace capacitor C4, a trace capacitor C S , a linearity inductor L LIN , and a linearity resistor R 5 , that are coupled in a well-known manner. A supply voltage B+ is coupled to circuit 103 via a winding T2a of a transformer T2. During retrace, circuitry 103 forms a retrace resonant circuit. FIGURES 2a-2f and 3a-3f illustrate waveforms useful for explaining the operation of the circuit of FIGURE 1. Similar symbols and numerals in FIGURES 1 and 2a-2f and 3a-3f indicate similar items or functions. In operation, driver transistor Q1 of FIGURE 1 is conductive as a result of a signal V OSC of FIGURE 2a being positive until time t 1 of FIGURE 2c that causes transistor Q2 to be nonconductive. The magnetic energy supplied by a collector current I c1 of transistor Q1 is stored in winding W1 of FIGURE 1. When transistor Q1 is turned off, during interval t 1 -t 3 of FIGURE 2a, the stored energy produces in winding W2 of FIGURE 1 a forward base current I b2 of transistor Q2. Current I b2 is sufficient to turn on deflection transistor Q2 and to maintain it in saturation prior to time t 2 of FIGURE 2e. The amplitude of current I b2 of FIGURE 2d remains substantially constant during the first half of trace, interval t 1 -t 2 . This is so because windings W2 and W3 are short circuited by the low impedance of the base-collector junction of transistor Q2 that is coupled in series with the forward biased damper diode D2. At time t 2 , a deflection current I H in deflection winding L H of FIGURE 1 reversed polarity. As a result of current reversals in deflection winding L H of circuitry 103 of FIGURE 1 and in a fly back transformer winding T2a collector current I c2 starts flowing in transistor Q2 of FIGURE 1 after time t 2 of FIGURE 2e in an upramping manner. Upramping collector current I c2 that flows as an emitter current through winding W3 of FIGURE 1 is transformer coupled through current transformer T1 to the base of transistor Q2 in accordance with a turns ratio N2:N3 of windings W2 and W3, respectively. Transformer coupled current I c2 provides a major portion of upramping forward base current I b2 of transistor Q2, during the second half of trace, interval t 2 -t 3 , when collector current I c2 is upramping. The base-emitter junction of transistor Q2 forms a very low impedance across winding W2. Therefore, a performance similar to that of an ideal current transformer operation is obtained in transformer T1 for coupling the current in winding W3 to winding W2. The required turns ratio is selected in accordance with a current gain parameter h FE of transistor Q2. For example, when parameter h FE is equal to 4, the selected turns ratio, N2:N3, may be equal to 4. The turns ratio between windings W2 and W3 is determined by the current gain h FE of transistor Q2. Should the core selected for transformer T1 provide tight magnetic coupling, such as for example, when an E-type core is utilized, such turns ratio may be equal to the value of current gain h FE of transistor Q2. On the other hand, if the magnetic coupling is not tight, a lower turn ratio may be chosen. Upramping base current that is proportional to current I c2 is induced in winding W2 of FIGURE 1 only after time t 2 of FIGURE 2d. Transformer coupled upramping current I c2 of FIGURE 2e is added to the aforementioned current already provided by the stored magnetic energy in transformer T1 to form upramping base current I b2 . The rate of change of current I b2 of FIGURE 1, advantageously, closely tracks the base current requirement of deflection transistor Q2, having upramping collector current I c2 , in a manner that maintains transistor Q2 in saturation without significantly overdriving the base of transistor Q2, during the second half of trace interval, t 2 -t 3 . Cut-off of transistor Q2 of FIGURE 1 is initiated at time t 3 of FIGURE 2e, when transistor Q1 of FIGURE 1 is switched into saturation. The level of collector voltage V c1 of FIGURE 2b at time t 3 and the turns ratio between windings W1 and W2 of FIGURE 1 determine the amplitude of the negative turn-off voltage developed across the base-emitter junction of transistor Q2. This voltage and the leakage inductance of transformer T1 determine the negative slope dI b2 /dt of FIGURE 2d during a storage or turn-off delay interval t 3 -t 4 of transistor Q2 of FIGURE 1. By selecting a proper turns ratio of windings W1 and W2, that is, for example, between 5 and 10, the desired negative slope dI b2 /dt for optimizing the base carrier discharge can be obtained. This is so because the turns ratio between windings W1 and W2 of transformer T1 determines the leakage inductance of transformer T1. The waveform of collector current I c1 of FIGURE 2c shows the peak negative base current used for turning off transistor Q2 at time t 4 of FIGURE 2c, the start of horizontal retrace. During a substantial portion of the second half of trace, interval t 2 -t 3 of FIGURE 2d, the increasing forward base current causes an increasing base-emitter voltage of transistor Q2 of FIGURE 1, which is transformer coupled across winding W3. This transformer coupled voltage is coupled across winding W3 and is coupled in series with deflection transistor Q2. Such series coupled voltage, however, is negligibly small. Therefore, the voltage across winding W3 does not appreciably impair the linearity of the deflection current and of current I c2 of FIGURE 2c. Advantageously, resistor R3 of FIGURE 1 serves to supply a base current that drives transistor Q1 into saturation immediately after oscillator 102 is turned off or made inoperative. Oscillator 102 may be turned off in response to an output signal ON/OFF of a remote receiver, not shown, that is used for turning off the television receiver to operate the receiver in a stand-by mode operation. Advantageously, by being conductive, transistor Q1 prevents self-oscillation of deflection circuit output stage 101 during the transition to standby mode operation. Such self oscillation might have occurred in the absence of drive pulses from the oscillator 102 and might have caused harmful high current and voltage peaks in output stage 101. Such high peaks might have occurred if driver transistor Q1 were not held in conduction as long as energy is circulating in deflection circuit output stage 101 following the transition to standby mode. The stored energy in the driver transformer might have caused output transistor Q2 to turn on, but not to turn off properly . Transistor Q2 might not turn off properly because the negative going slope of base current I b2 , that in normal operation causes transistor Q2 to turn off, is not produced after oscillator 102 ceases operation. By maintaining transistor Q1 conductive during the transition to standby mode of operation, any lengthening of the last deflection cycle, occurring after oscillator 102 ceases operation, is avoided. Such lengthening, disadvantageously, might have overstressed deflection transistor Q2. Resistor R1 limits the collector current and power dissipation in transistor Q1, during the transition to stand-by mode of operation. The use of a rather high value for resistor R1 such as, for example, 180 ohm is, advantageously, possible. This is so because driver transistor Q1 operates at a lower power level than the analogous transistor in the aforementioned prior art circuit. Consequently, the collector current in transistor Q1, that is in saturation during a transition to standby mode, will not harm transistor Q1. Because resistor R1 has a high value, collector current and power dissipation in transistor Q1 are limited, in the case of a failure in oscillator 102. Advantageously, a main portion of forward base current I b2 of deflection transistor Q2 is derived from its collector current I c2 and rises linearly. In contrast, in the aforementioned prior art driver circuit, the entire drive current is supplied by the energy stored in the driver transformer. Therefore, a lower dissipation occurs in driver stage 100 of FIGURE 1. The result is that the required level of voltage V+ may be, advantageously, lower than in such prior art driver circuit. Controlling the wave shape and peak amplitude of base current I b2 in a precise manner is desirable to insure saturation of deflection transistor Q2 during the second half of trace. This feature, advantageously, is achieved in the circuit of FIGURE 1 by transformer coupling the emitter current of transistor Q2 that is linearly upramping via winding W3. In comparison, in the aforementioned prior art circuit, an R-C damping network coupled across a primary winding of the coupling transformer is typically used for controlling the waveform of the drive current. However, tolerances of the R-C damping network produce, disadvantageously, a corresponding tolerance spread in the values of the drive current. FIGURES 3a an 3c illustrate waveforms of signal V OSC of FIGURE 1 having different duty cycles and the same frequency and the resulting waveforms of base current I b2 of FIGURES 3b and 3f, respectively. As can be seen, the peak amplitude of base current I b2 is the same in both cases. The collector current I c2 of FIGURE 3c is also unaffected by the duty cycle variation of signal V OSC . FIGURE 4 illustrates a deflection circuit for a pin-free 90° CRT that includes a driver stage similar to that shown in FIGURE 1. The corresponding waveforms are shown in FIGURES 7a-7d. Similarly, FIGURE 5 illustrates a deflection circuit for a 110° CRT using a diode modulator. The corresponding waveforms are shown in FIGURES 2a-2f and 3a-3f. FIGURE 6 illustrates a deflection circuit for a 110° CRT using a switched East-West raster correction circuit. The applicable waveforms are shown in FIGURES 7a-7d. Similar symbols and numerals in FIGURES 1, 2a-2f, 3a-3f, 4-6 and 7a-7d indicate similar items or functions. The reason that the waveforms of FIGURES 7b and 7c are different from those in FIGURES 2d and 2e, respectively, and from those in FIGURES 3b and 3c, respectively, is that in FIGURES 4 and 6 the damper diode is coupled across transistor Q2. The damper diode of FIGURE 4, for example, forms with transistor Q2 an integrated circuit. This results in the deflection current in winding L H to flow also through winding W3, during the first half of trace. In contrast, in FIGURE 1, deflection current I H does not flow through winding W3 during the first half of trace because the cathode of diode D2 is coupled to ground directly and not via winding W3. Transformer T1 of FIGURE 6, may, preferably, have a lower primary winding inductance and a slightly higher turns ratio between windings W2 and W3 than in the circuit of FIGURE 1 having a discrete damper diode D2. In an arrangement where initiating standby mode of operation is obtained by switching off voltage V+ of FIGURE 1, resistor R3 may, preferably, be coupled to a positive voltage V+', as shown in broken line. Voltage V+' is derived from a secondary winding T2b of a flyback transformer T2, rectified by a diode 77 and filtered by a capacitor 78. This will ensure a fail safe transition from normal to standby operation by maintaining transistor Q1 in saturation during the aforementioned transition to standby mode operation. FIGURE 8 illustrates a deflection circuit for a CRT operating at a deflection frequency of 2xf H , that is approximately 32 KHZ. Similar symbols and numerals in FIGURES 1 and 8 indicate similar items or functions. The driver stage of FIGURE 8 is similar to that shown in FIGURE 1. As shown in FIGURES 1, 4-6 and 8, driver stage 100 can be easily adapted to many types of horizontal output transistors and to different yoke impedances and is well suited for higher scan rate applications. The component count required for producing the upramping base current is reduced relative to that required in some prior art circuits. The upramping base current improves the efficiency of the driver stage. The well controlled base driver waveform produces a lower dissipation in the output stage and an increased reliability.

Claims: A television deflection apparatus, comprising: a deflection retrace resonant circuit (C 4 ,C S ,LIN,L H ) including a deflection winding (L H ) and a retrace capacitance (C 4 ); a first switching transistor (Q2) responsive to a control signal (I b2 ) at a frequency that is related to a deflection frequency and coupled to said retrace resonant circuit for performing a switching operation that generates a deflection current (I H ) in said deflection winding (L H ); a transformer (T1) having a secondary winding (W2) that is coupled to a control terminal of said first switching transistor (Q2) for developing said control signal at said control terminal; a source (102) of an input signal (V OSC ) at a frequency that is related to a deflection frequency; and a second switching transistor (Q1) responsive to said input signal and coupled to a primary winding (W1) of said transformer (T1) for generating a first current (I C1 ) in said second switching transistor and in said primary winding of said transformer that is transformer coupled via said transformer to said control terminal of said first switching transistor (Q2) to develop said control signal at said control terminal during normal operation, characterized by means (R3,T2b) responsive to an on/off control signal, which turns off said source (102) of said input signal (V OSC ) to initiate a standby mode of operation in which said second transistor (Q1) is turned off, for maintaining said second switching transistor (Q1) conductive during a transition interval that precedes a change from normal operation to said standby mode of operation, such that said first current (I C1 ) flows in said primary winding and in said second switching transistor, during said transition interval.

Inventors: HENNIG BRUNO EMANUEL, HENNIG, BRUNO EMANUEL
Applicants: RCA THOMSON LICENSING CORP, RCA THOMSON LICENSING CORPORATION
Application year: 1990
Main classification: H03K4
IPC codes: H03K4, H04N3