Switching power supply device

Abstract: An improved regenerative control type power supply device having a higher efficiency is disclosed. The power supply device has: a first switching elements which is connected in series with the primary winding of a transformer (this series circuit is connected to the terminals of a DC power source); a second switching element; a capacitor which is connected in series with the second switching means (this series circuit is connected to the primary winding); a rectifying and smoothing circuit for rectifying and smoothing a flyback voltage which generates in the secondary winding and for outputting a resulting DC voltage; and a drive circuit for driving alternately the first and second switching means with predetermined ON-OFF periods.

Description: The present invention relates to a first switching means connected in series with the primary winding of the transformer, this series circuit being connected to the terminals of the DC power source, a second switching means, a capacitor connected in series with the second switching means, the series circuit including said capacitor and said second switching means is connected across the primary winding, a rectifying and smoothing circuit, and a drive means for driving alternately said first and second switching means with predetermined ON-OFF periods, said drive means is formed to respond to the DC output voltage from said rectifying and smoothing circuit. Such a switching power supply device is known from US-A-4 809 148. This known switching power supply device is formed as a forward converter. The second controllable switch and the capacitor of the converter are connected in parallel with the first controllable switch and the first and second controllable switches are complementary. With such a forward converter it is difficult to realize a zero-cross turn-on operation. Moreover with this known forward converter the winding voltage is liable to be clamped to 0 V after the turn-off of the second switch and the charges stored in the capacitors cannot be sufficiently regenerated, but are discharged through the first switch. The result of this is a turn-on loss at the first switch. From DE-A-3 840 385 a current supply device is known which includes at least three switching elements within a primary circuit, one switch element being used as an interruptor switch, whereas the remaining two switches are part of an inverter circuit. From US-A-4 729 085 a resonant regulator power supply is known which comprises a resonant circuit coupled to a source of alternating input that is controllable in operating frequency. A power transformer primary winding is coupled across the capacitive element of the resonant circuit for generating an output voltage across a secondary winding. The operating frequency can be varied by a control circuit in a negative feedback loop to regulate the output voltage. Among conventional switching power supply devices, a self-excited flyback type switching power supply device has been widely used, because it consists of fewer parts and can be manufactured at a relatively low cost. It is known, however, that the switching frequency of such a device fluctuates depending upon the degree of the output current, with the result that interferences occur in the operation of an associated electronic apparatus and that a larger size rectifying and smoothing circuit is required. In an attempt to overcome such problems with the prior art devices, a switching power supply device of the regenerative control type has been proposed. Figure 7 shows a switching power supply device of such a type. The device of Figure 7 comprises a DC power source 1, a transformer 3, switching elements 4 and 14, a diode 5, a synchronizing oscillation circuit 6, a rectifying diode 7, a smoothing capacitor 8, a secondary switching element 14, and a control circuit 15. The DC power source 1 rectifies an AC voltage and smoothes the resulting DC voltage. Alternatively it may consist of a battery or the like. The positive and negative terminals of the DC power source 1 are connected to input terminals 2 and 2', respectively. The transformer 3 has: a primary winding 3a connected at one terminal to the input terminal 2' through the switching element 4; a secondary winding 3c connected at one terminal to an output terminal 10' and at the other terminal to an output terminal 10 through the rectifying diode 7; and a bias winding 3b connected at one terminal to the input terminal 2' and at the other terminal to the synchronizing oscillation circuit 6. The switching element 4 turns ON or OFF in response to ON/OFF signals which are supplied at the control terminal from the synchronizing oscillation circuit 6, thereby applying the input voltage to the primary winding 3a and interrupting the voltage. The synchronizing oscillation circuit 6 actuates the switching element 4 to turn ON and OFF periods of time. The OFF period continues until the polarity of the induced voltage of the bias winding 3b is inverted. This repeated ON/OFF cycle causes continuous oscillation. The energy accumulated in the transformer 3 during an ON period of the switching element 4 is released from the secondary winding 3c through the rectifying diode 7 or the switching element 14 to the smoothing capacitor 8 during an OFF period of the switching element 4. After the above-mentioned energy release, a secondary current is caused to flow reversely from the smoothing capacitor 8 to the secondary winding 3c through the switching element 14. The period of this reverse flow is controlled by the control circuit 15. The rectifier diode 7 is connected at its anode to one terminal of the secondary winding 3c and at its cathode to the output terminal 10. The smoothing capacitor 8 is connected between the output terminals 10 and 10'. The induced voltage appearing across the secondary winding 3c is rectified by the rectifying diode 7, and then smoothed by the smoothing capacitor 8 to provide an output voltage. The control circuit 15 compares the voltage appearing across the output terminals 10 and 10' with an internal reference voltage in order to vary the above-mentioned flow period of the secondary current through the secondary switching element 14. The operation of the power supply device is described with reference to Fig. 8. In Fig. 8, (a) shows the waveform of a voltage V DS appearing across the switching element 4; (b) the primary current I D flowing through the primary winding 3a; (c) the waveform of driving pulse V G1 output from the synchronizing oscillation circuit 6; (d) the secondary current I D flowing through the secondary winding 3c; and (e) the waveform of a driving pulse V G2 for the secondary switching element 14. The hatched portions in (e) of Figure 8 indicate reverse flow periods for causing the secondary current I o to reversely flow in the secondary winding 3c. A magnetic flux develops in the transformer 3 as the primary current I D flows through the primary winding 3a during an ON period of the switching element 4 which period is determined by the synchronizing oscillation circuit 6, so that energy is accumulated in the transformer 3, whereupon an induced voltage develops in the secondary winding 3c. It is so arranged that the rectifying diode 7 is reversely biased by the inuced voltage and that the switching element 14 remains in its OFF-position. When the switching element 4 turns OFF in response to an OFF signal from the synchronizing oscillation circuit 6, a flyback voltage develops in the primary winding 3a, and, simultaneously, a flyback voltage is induced in the secondary winding 3c in such a direction that the rectifying diode 7 is forward biased. Accordingly, the energy accumulated in the transformer 3 is released as a secondary current I o through the secondary winding 3c, which is then smoothed by the smoothing capacitor 8 and supplied as an output voltage to the output terminals 10 and 10'. In this case, the switching element 14 is actuated to turn ON by the control circuit 15, but there occurs no particular operation change, through whichever the diode 7 or the switching element 14 the secondary current may flow. When all the energy accumulated in the transformer 3 is released until the secondary current becomes zero, the voltage appearing across the smoothing capacitor 8, that is, output voltage, is applied to the secondary winding 3c through the switching element 14 which has been already in ON state, and accordingly the secondary current flows reversely from the smoothing capacitor 8, so that a magnetic flux is generated in the reverse direction in the transformer 3, thereby causing energy to be accumulated therein. In this condition, there is no change in the polarity of the induced voltage developing in each winding of the transformer 3, therefore, there is no change in the flyback voltage in the bias winding 3b. Accordingly, the synchronizing oscillation circuit 6 causes the switching element 4 to remain in the OFF state. As mentioned above, the ON period of the switching element 14 is controlled by the control circuit 15. When the switching element 14 becomes OFF, the induced voltage in each winding of the transformer 3 is inverted in polarity. Therefore, the induced voltage developing in the secondary winding 3c causes the rectifying diode 7 to be reversely biased. Since the switching element 14 is in the OFF state, the secondary winding current does not flow. In the primary winding 3a, the induced voltage develops in such a direction that the voltage at the terminal to which the switching element 4 is connected is negative, and, on the other hand, the voltage at the terminal to which input terminal 2 is connected is positive. Therefore, the primary current flows in such a direction that the DC power source 1 is charged through the diode 5, so that the energy accumulated in the transformer 3 during the OFF period is supplied to the DC power source 1 (i.e., the power regeneration is conducted). At this time, the polarity of the induced voltage developing in the bias winding 3b is also inverted, and accordingly the synchronizing oscillation circuit 6 actuates the switching element 4 to turn ON. In this case, there is no particular operational change, through whichever the diode 5 or the switching element 4 the primary current may flow. When all the energy accumulated in the transformer 3 during the OFF period is released until the primary current is reduced to zero, the primary current flows from the DC power source 1 through the switching element 4 which has been already in the ON state, so that the transformer 3 is charged in the direction contrary to the that of the above-mentioned discharge, with the result that a magnetic flux develops in the transformer 3 and energy is thus accumulated therein. In this condition, there is no change in the polarity of the induced voltages developing in each windings of the transformer 3, and accordingly the synchronizing oscillation circuit 6 maintains to control the switching element 4 to be kept in the ON state. When the switching element 4, the ON period of which is determined by the synchronizing oscillation circuit 6, is actuated to turn OFF, the energy accumulated in the transformer 3 is released as the secondary current through the secondary winding 3c. Cycles of these operations are repeated so that the output voltage is continuously supplied across the output terminals 10 and 10'. The manner of performing the steady control of the output voltage will be described. In Figure 8 showing the waveforms at various portions of the power supply device of Figure 7, the OFF period (between times t 1 and t 3 ) of the driving pulse V G1 in the synchronizing oscillation circuit 6 is represented by T OFF , the reverse flow period (between times t 2 and t 3 ) of the secondary current I o is represented by T' OFF , the ON period (between times t 3 and t 5 ) is represented by T ON , and the regenerative period (between times t 3 and t 4 ) of the primary current I D is represented by T' ON . Then, the current I OUT output from the output terminals 10 and 10' can be expressed by: I OUT = 1 2 x V OUT x 1 L S x T OFF T x (T OFF - 2T' OFF ) The output voltage V OUT can be expressed by: V OUT = V IN x N S N P x T ON T OFF = V IN x N S N P x T' On T' OFF = V IN x N S N P x T ON - T' ON T OFF - T' OFF The oscillation frequency f is expressed by; f = 1 T ON + T OFF = 1 T In the above expressions, N S represents the number of turns of the secondary winding 3c; N p represents the number of turns of the primary winding 3a; L s represents the inductance of the secondary winding 3c; and V IN represents the input voltage supplied from the DC power source 1. The ON period T ON is kept at a constant value determined by the synchronizing oscillation circuit 6. If the output voltage V OUT is constant, therefore, the OFF period T OFF is constant, and the oscillation frequency f is also constant. However, the reverse flow period T' OFF may be varied by the secondary switching element 14 controlled by the control circuit 15, and K=(1/2) · V OUT · (1/L S ) · (T OFF /T) in the expression (1) is constant when the output voltage V t isconstant. Even if the output current I OUT is varied, therefore, it can be controlled by changing the reverse flow period T' OFF . Even if the input voltage V IN is varied, moreover, it can be controlled by changing the reverse flow period T' OFF' as seen from the expression (2). Therefore, the output voltage V OUT can be controlled so as to be always kept constant by changing the reverse flow period T' OFF . The reverse flow period T' OFF can be changed by controlling the ON period of the switching element 14 which is controlled by the control circuit 15. Figure 9 shows various waveforms obtained when the output current I OUT is changed. In Figure 9, solid lines indicate waveforms obtained when the output current I OUT flows at the maximum level from the output terminals 10 and 10', or at the so-called maximum load period, and broken lines indicate waveforms obtained when the output current I OUT is zero. or at the so-called lo-load period. When the input voltage is constant, the ON Period T ON is constant, and accordingly flux variation range ΔB is always constant. In such a regenerative control type switching power supply device, when the switching element 4 turns OFF, a surge voltage occurs due to the leakage inductance in the transformer 3. At the maximum load, the level of the surge voltage is approximately the same extend as in a conventional self-excited flyback type switching power supply device, and, at a light load, the level of the surge voltage is greater than that caused in such a self-excited flyback type power supply device because the peak value of primary current is considerably higher immediately before the turn OFF. The conventional regenerative control type switching power supply device provides an advantage that because of its ability to regenerate energy at the turn ON of the switching element 4, even when a snubbing capacitor is connected between the both terminals of the switching element 4, the surge voltage at the turn-on can be efficiently restrained without involving any turn-on loss. However, the resonance energy due to the capacitor and the leakage inductance of the transformer 3 is considerably great, so that ringing waveforms are superposed over another throughout each OFF period, which becomes a source of noise. The addition of a larger snubbing capacitor becomes a greater hindrance to realization of a higher switching frequency for compactness of the power source. When the input DC power source is obtained from an AC power source through a rectifying and smoothing circuit, it is usual that the rectifying and smoothing circuit is of the capacitor input type which comprise a smoothing element and a capacitor and that the capacitor is used as an input capacitor which serves as a DC power source. Generally, it is desired that power supply devices have a higher efficiency and be small in size, and on the other hand the output holding time is required to be set in order to protect electronic apparatuses as a load from possible troubles such as momentary interruption of input power. In the prior art arrangement, the output holding time depends largely upon the static capacitance of the input capacitor, and, therefore, the static capacitance is determined by both the power capacity of the power source and the output holding time. For this reason, even when there is a sufficient ripple withstand capacity, there are cases where use of a larger input capacitor is required. In addition, the conducting period of the input current from the AC power source is shorter in the stage of steady operation so that the peak value of the input current becomes larger, thereby causing the problem in that the power factor and efficiency substantially drop. The proceedings of the Annual Applied Power Electronics Conference and Exposition (APEC), March 13-17, 1989, IEEE, p. 413 to 422, "Development of a DC-To-DC Power Converter For Distributed Power Processing" issued by Lo at all ., refers to a switch-mode, interleaved flyback power converter including a circuit according to which the switch duty ratio or the power switch on-time is varied to provide output voltage regulation. DE-A- 36 17 374 discloses a power converter which includes circuity to switch the primary circuit on the basis of a comparison of a reference voltage with a voltage across the primary winding, on the one hand, and, on the other hand, on the basis of a voltage derived from the secondary winding circuit in combination with a reference voltage and a voltage being proportional with a signal from the primary circuit. It is an object of the present invention to provide a switching power supply device which can efficiently inhibit any possible surge voltage and/or ringing waveform at the turn-OFF of the switching element, without any detriment to the advantages characteristics of the regenerative control type switching power supply device such that variations in switching frequency due to changes in load can be suppressed. Moreover the switching power supply device according to the present invention should also be able to effectively reduce any possible switching noise and to reduce the static capacitance of the input capacitor without shortening the output holding time and thus broadening the conduction period of the input current. According to the present invention this object is solved by the subject-matter of claim 1. According to the present invention the effect of extending the output holding time is attained and moreover the switching power supply device of the present invention can effect the zero-cross turn-ON operation. Improved embodiments of the inventive switching power supply device are disclosed within the subclaims 2 to 4. Thus, the invention described herein makes possible the objectives of: (1) providing a switching power supply device which can efficiently inhibit any possible surge voltage and/or ringing waveform at the turn-off of the switching element, without detriment to the advantageous characteristics of the regenerative control type switching power supply device such that variations in switching frequency due to changes in load can be suppressed; (2) providing a switching power supply device which can effectively reduce any possible switching noise; and (3) providing a switching power supply device which can reduce the static capacitance of the input capacitor without shortening the output holding time and thus broaden the conduction period of the input current. BRIEF DESCRIPTION OF THE DRAWINGS This invention may be better understood and its numerous objects and advantages will become apparent to those skilled in the art by reference to the accompanying drawings as follows: Figure 1 is a circuit diagram of a power supply device according to the invention. Figure 2 shows waveforms at various portions of the device of Figure 1. Figure 3 is another circuit diagram of a power supply device according to the invention. Figure 4 is a further circuit diagram of a power supply device according to the invention. Figure 5 is a graph showing the variation of the input and output voltages of the device of Figure 4. Figure 6 shows waveforms of the input voltage and the input current of the device of Figure 4. Figure 7 is a circuit diagram of a conventional power supply device. Figures 8 and 9 show waveforms of various portions of the device of Figure 7. DESCRIPTION OF THE PREFERRED EMBODIMENTS According to the invention, even if the potential of the first switching means is raised as a result of the turn-off of the switching means, the potential is clamped by the second switching means and the capacitor, and accordingly the occurrence of any surge voltage is inhibited. In addition, since the second switching means is ON at that time, the voltage present across the capacitor is applied to the primary winding of the transformer, there being little or no possibility of ringing occurrence. According to the invention, the resonance capacitor connected to the switching means or to the both terminals of the switching means resonates with the leakage inductance of the transformer during the turn-on or turn-off operation, thereby lowering the rate of variation of the operating voltage, so that the level of switching noise can be reduced. Moreover, the energy accumulated in the capacitor connected in series with the second switching means to the terminals of the primary winding of the transformer is released from the secondary winding through the transformer during each ON period of the second switching means, and therefore, longer output holding time can be obtained as compared with the prior art arrangement. Figure 1 shows a switching power supply device according to the invention. In Figure 1, elements equivalent to those in Figure 7 are designated by like reference numerals, and their detailed description is omitted. The device of Figure 1 comprises a DC power source 1, a transformer 3, a switching element 4, a diode 5, a rectifying diode 7, a synchronising oscillation circuit 6, a smoothing capacitor 8, and a control circuit 9. The switching element 4 and the diode 5 are connected in parallel, and constitute the first switching means. The switching element 4 is controlled by the synchronizing oscillation circuit 6. The rectifying diode 7 and the smoothing capacitor 8 constitute the rectifying and smoothing circuit. The transformer 3 comprises a primary winding 3a, a secondary winding 3c, and a bias winding 3b. The device of Figure 1 further comprises another switching element 11 which is controlled by the control circuit 9, another diode 12, and a capacitor 13. The switching element 11 and the diode 12 are connected in parallel, and constitute the second switching means. The control circuit 9 has input portions connected respectively to output terminals 10 and 10'. The input portions of the control circuit 9 are electrically separated from the output portions connected to the switching element 11. The operation of the switching power supply device will be described with reference to Figure 2 in which waveforms at various portions of the device are shown. In Figure 2, (a) shows the waveform of a voltage V DS present across the switching element 4; (b) a primary current I D flowing through the switching element 4 or the diode 5; (c) the waveform of a driving pulse V G1 output from the synchronizing oscillation circuit 6; (d) a primary current I c flowing through the switching element 11 or the diode 12; (e) the waveform of a driving pulse V G2 for the switching element 11; (f) a secondary current I o flowing through the secondary winding 3c; and (g) the variation in the magnetic flux  of the transformer 3. As the primary current I D flows through the primary winding 3a during an ON period of the switching element 4 which period is determined by the synchronizing oscillation circuit 6, a magnetic flux develops in the transformer 3 so that energy is accumulated in the transformer 3, whereupon an induced voltage develops in the secondary winding 3c. The secondary winding 3c is wound in such a manner that the rectifying diode 7 is reversely biased by this induced voltage. It is so arranged that the diode 12 in the primary side is reversely biased by this induced voltage, and also that the switching element 11 is controlled so as to remain in its OFF-position under this condition. When the switching element 4 turns OFF in response to an OFF signal from the synchronizing oscillation circuit 6, a flyback voltage develops in the primary winding 3a, and the diode 12 is forward biased. Simultaneously, a flyback voltage is induced also in the secondary winding 3c in such a direction that the rectifying diode 7 is forward biased. Accordingly, the energy accumulated in the transformer 3 is released as a primary current I c through the primary winding 3a and the diode 12 and also as a secondary current I o through the secondary winding 3c. The primary current I c is then smoothed by the capacitor 13 to be supplied as a DC voltage V c , and the secondary current I o is smoothed by the smoothing capacitor 8 to be supplied as an output voltage to the output terminals 10 and 10'. In this case, the switching element 11 is actuated to turn ON by the control circuit 9, but there occurs no particular operation change, through whichever the diode 12 or the switching element 11 the primary current I c may flow. When capacitance components such as a parasitic capacitance are not considered, the energy accumulated in the transformer 3 begins to be released first from the primary winding 3a under the influence of the leakage inductance, at the voltage reverse in each windings of the transformer 3 caused by the OFF operation of the switching element 4. That is, the primary current I c begins to flow with the initial value which coincides with the final value I p of the primary current current I D , while the secondary current I o rises from zero. In this case, the magnetic flux  in the transformer 3 decreases linearly, because the accumulated energy in the transformer 3 is released while the DC voltage V c is applied to the primary winding 3a. Accordingly, the primary current I c decreases monotonously until it is finally reduced to 0 Amp. Thereafter, since the switching element 11 is in the ON state, the discharge current from the capacitor 13 begins to flow to the primary winding 3a through the switching element 11. Since the DC voltage V c is applied to the primary winding 3a, the rectifying diode 7 is forward biased, so that the secondary current I o continues to flow. After the energy accumulated in the transformer 3 has been completely released while the switching element 4 is ON, the DC voltage V c remains to be applied by the switching element 11, whereby the transformer 3 is reversely excited and energy is accumulated therein in the inverse direction. When the switching element 11 is turned OFF by the control circuit 9, the voltages of the windings of the transformer 3 are inverted, and the rectifying diode 7 is reverse-biased, so that the flow of secondary current I o is ceased. Thus, the induced voltage in the primary winding 3a has a direction such that the polarity at the terminal to which the switching element 4 is connected is negative and the polarity at the terminal to which the input terminal 2 is connected is made positive. Accordingly, primary current I D flows through the diode 5 in a direction that the DC power source 1 is charged, whereby the energy accumulated in the transformer 3 during the OFF period is supplied to the DC power source 1 for power regeneration. In this case, the polarity of the induced voltage developing in the bias winding 3b is also inverted, and, accordingly, the synchronizing oscillation circuit 6 actuates the switching element 4 to turn ON; it is noted, however, that there is no particular change in operation, through whichever the switching element 4 or the diode 5 the primary current I D may flow. When all the energy accumulated in the transformer 3 during the OFF period is released until the primary current is reduced to zero, the primary current I D flows from the DC power source 1 through the switching element 4 which is already in the ON state, in the direction opposite to the direction of the discharge, so that magnetic flux develops in the transformer 3, thereby accumulating energy therein. In this condition, there is no change in the polarity of the induced voltage developing in each windings of the transformer 3, and the switching element 4 is kept in the ON state by the synchronizing oscillation circuit 6. When the switching element 4 operating in ON intervals as determined by the synchronizing oscillation circuit 6 is turned OFF, the energy accumulated in the transformer 3 is released through the primary winding 3a to the capacitor 13, and is also released as the secondary current I o through the secondary winding 3c to the output. Through repetition of cycles of these operations the output voltage is continuously supplied from the output terminals 10 and 10'. The manner of performing the steady control of the output voltage in the device of Figure 1 will be described. In Figure 2 showing the waveforms at various portions of the power supply device of Figure 1, the OFF period (between times t 1 and t 3 ) of the driving pulse V G1 in the synchronizing oscillation circuit 6 is represented by T OFF , the period (between times t 2 and t 3 ) of reverse-exciting the transformer 3 is represented by T' OFF , the ON period (between times t 3 and t 5 ) is represented by T ON , and the regenerative period (between times t 3 and t 4 ) of the primary current I D is represented by T' ON . During the steady operation of the switching power supply device of the embodiment, there is little or no fluctuation in the DC voltage V c because the capacity of the capacitor 13 is sufficiently large. The primary current I c in the OFF period which is a ripple current is equal in both the charging and discharge periods, with the average current of 0 Amp. Therefore, the energy which is emitted from the secondary winding 3c and supplied from the output terminals 10 and 10' is equal to the difference between the energy accumulated in the transformer 3 during the ON period and the energy supplied to the DC power source 1 for regeneration during the period T' ON . The DC voltage V c is expressed by the following expression: V c = T ON T OFF X V IN = T' ON T' OFF X V IN This becomes when, in the operation for stabilization of the output voltage as already described with respect to the prior art regenerative control type switching power supply device, the DC voltage V c is considered to be a no-load output voltage. Further, from the fact that the output voltage V OUT in the switching power supply device of the embodiment is obtained by rectifying the flyback voltage of the secondary winding 3c, the following relation is obtained: Thus, it will be found that the output voltage V OUT can be regulated by adjusting the DC voltage V c . For example, when the output current I OUT has been decreased and in turn the output voltage V OUT has risen, the ON period of the switching element 11 (ie., the OFF period TOFF of the switching element 4) is made longer by the control circuit 9, so the amount of outgoing charge in the capacitor 13 becomes greater than the amount of incoming charge, with the result that the DC voltage V C is reduced. As the DC voltage V C decreases, the output voltage V OUT also decreases, and the voltage V C developed in and applied to the windings of the transformer 3 drops. Therefore, the gradient of the primary current I c is moderated so that the DC voltage V C finally reaches a level at which the output voltage V OUT becomes a predetermined value. That is, the output voltage V OUT can be stabilized by regulating the ON period of the switching element 11. primarily, there is no much room for variation of the DC voltage V C for the purpose of correcting the fluctuation (load regulation) of the output voltage V OUT due to the fluctuation of the output current I OUT . If the ON period T ON is constant, therefore, the OFF period T OFF is little liable to variation, end the switching frequency and magnetic flux variation range ΔB are generally constant as well. General aspects in this regard are shown by broken lines in Figure 2. Another embodiment of the invention will described with reference to Figure 3. The embodiment of Figure 3 has a similar construction as that of the embodiment of Figure 1 except that a capacitor 16 is connected in parallel with diode 5 and another capacitor 17 is disposed so as to bridge the terminals of the secondary winding 3c. The capacitors 16 and 17 function as resonance capacitors. By connecting the resonance capacitor 16 to the switching element 4 as shown in Figure 3, it is possible to moderate possible rapid rise of the voltage waveform at each turn-off of the switching element 4. The switching power supply device of the embodiment has a similar power regeneration as that of the afore-described prior art regenerative type switching power supply device. The electric charge accumulated in the resonance capacitor 16 is supplied to the DC power source 1 for power regeneration during the turn-off period of the switching element 11, and therefore it is not reduced to a turn-on loss at the switching element 4. The resonance capacitor 17 connected to both terminals of the secondary wiring 3c can suppress a surge voltage which may develop across the rectifying diode 7 with the steep decrease to 0 Amp. of the secondary current I o at the turn-off of the switching element 11. Operations other than those in transition stages are the same as those before described with respect to the embodiment of Figure 1, and therefore their description is omitted. The addition of these resonance capacitors has the effect that in transition stages, there will occur changes in the output impedance of each windings of the transformer 3 and, more particularly, during the OFF period of the switching element 4 there will be some change in the initial value of the current in each windings. However, this little affects the control operation, and rather produces an advantage in that the peak value of the primary current I c during the OFF period is lowered so that current loss is lowered as a whole. The resonance capacitor 16 may be connected to both terminals of the primary windings 3a or both terminals of the switching element 11. In either case, similar excellent effects can be obtained as well. The resonance capacitor 17 may be connected to both terminals of the rectifying diode 7, in which case similar effects can be obtained as well. In the above-described embodiments, the ON period T ON is fixed, and the output voltage V OUT is stabilized by regulating the OFF period T OFF . Alternatively, the OFF period T OFF is fixed, and the output voltage V OUT is stabilized by regulating the ON period T ON . Further, it is also possible that the switching cycle (T ON + T OFF ) of both the switching elements 4 and 11 is fixed, and the output voltage V OUT is stabilized by regulating the ON/OFF ratio. In any of the above mentioned ways, the invention can be carried out with same effect. Figure 4 shows a further switching power supply device according to the invention. The embodiment of Figure 4 comprises a full-wave rectifier circuit 19, an input capacitor 20 which is connected to the outputs of the full-wave rectifier circuit 19, and a control drive circuit 21. The embodiment a Figure 4 is so constructed that the DC power source can be obtained by rectifying and smoothing AC power, and the other components of this embodiment may be constructed in the similar manner with those of the embodiment of Figure 1. Namely, if the input capacitor 20 is taken as the DC input power source 1 in Figure 1, the manner of the operation of the third embodiment is the same as that in the first embodiment. The control drive circuit 21 may be constructed so that it includes the bias winding 3b of the transformer 3, the synchronizing oscillation circuit 6, and the control circuit 9 in Figure 1, or alternatively it may be a circuit which can execute such other output voltage stabilizing methods described above. Therefore, the description of the operation to this embodiment is omitted, and the manner of obtaining an output holding time which is longer as compared with that obtained in the conventional arrangement will be described with reference to Figure 5. In Figure 5, V o1 is a set value of the DC output voltage V OUT ; V o2 is the allowable lower limit of the DC output voltage V OUT ; V i1 is a value of the DC input voltage V IN during the steady operation; V i2 is the stabilized lower limit of the DC input voltage V IN at which the control drive circuit 21 outputs to the switching element 4 a driving pulse which has the maximum ON/OFF ratio; and V i3 is the halt voltage at which the control drive circuit 21 halts. The output holding time is a period of time from the stop of energy supply from the AC power source and until the DC output voltage V OUT drops to a level below the allowable lower limit V 02 . When the input is interrupted, the DC input voltage V IN begins to drop from the value V i1 , but until it reaches the stabilized lower limit V i2 the output voltage V OUT maintains the set value V o1 . During this period, the operation of the embodiment is similar to that of a conventional device. When the DC input voltage V IN further drops to be lower than the stabilized lower limit V i2 , the driving pulse from the control drive circuit 21 to the switching element 4 is fixed to the maximum ON/OFF ratio, and as the DC input voltage drops, the output voltage V OUT begins to drop from the set value V o1 . In the conventional switching power supply device, the output energy is supplied only from the input capacitor 20. In contrast, according to this embodiment, energy is also supplied from the capacitor 13 during the OFF period of the switching element 4, with the result that the DC input voltage V IN and output voltage V OUT decrease at the rates which are lower than those in the conventional arrangement. In Figure 5, curves indicated by "A" represent voltage variations in the present embodiment, and curves indicated by "B" represents those in the conventional arrangement. That condition continues until the DC input voltage V IN becomes lower than the halt voltage V i3 so that operation is stopped. During this course, the output voltage V OUT becomes lower than the allowable lower limit V o2 . In this way, according to the embodiment, the output holding time is longer than that in the conventional arrangement. According to the present embodiment, therefore, if the output holding time is to be maintained at the same level as in the prior art device, the static capacity of the input capacitor 20 can be decreased. Figure 6 is a waveform diagram showing waveforms of the DC input voltage V IN and input current I i in the case in which the static capacity of the input capacitor 20 is small (curves C), and the case in which the static capacity is large (curves D). As is apparent from Figure 6, if the input capacitor 20 which rectifies and smoothes the AC power source can be reduced in size, the conduction period of the input current I i can be lengthened so that the power factor can be improved. Moreover, the efficiency of the device can be improved by lowering the peak value of the input current I i . In the conventional regenerative control type switching power supply device, the rectifying element disposed at the secondary side of the transformer is provided with the switching element, and the energy at the secondary side is regenerated between the primary and secondary sides of the transformer thereby stabilizing the output. In contrast, according to the invention, a DC voltage obtained by rectifying and smoothing a flyback voltage in the primary winding is utilized, and the energy regeneration can be conducted using a rectifying element which is not provided with a switching means, whereby the output can be stabilized. Therefore, according to the invention, without detriment to the characteristic feature of the prior art arrangement in that there is no fluctuation in the switching frequency and the the magnetic flux variation range caused by the load, the voltage present during each OFF period of the primary side switching element is always clamped to the sum of the input voltage and the DC voltage, and it is thus possible to inhibit any excessive surge voltage and/or ringing during each turn-off period. Furthermore, the power regeneration is carried out on the primary side only and not between the primary side and the secondary side, so the power regeneration can be effected efficiently without producing loss of regenerated power due to the transformer efficiency. According to the invention, moreover, the regenerative capability of the device permits the zero-cross turn-on, end the addition of any resonance capacitor for reducing possible turn-off loss and/or high-frequency noise does not result in turn-on losses, but rather inhibits ringing occurrences as already mentioned, because the accumulated charge in the capacitor is supplied to the input power source for power regeneration. In particular, by the addition of such a resonance capacitor on the secondary side of the transformer, it is not only possible to protect the rectifying element against a surge voltage, but also possible to lower the peak value of the primary winding current during each OFF period. This naturally leads to an improved efficiency and improved reliability. Further, according to the invention, when a DC input power source is to be obtained from an AC power source through a rectifying circuit and an input capacitor, the dependence on the static capacity of the input capacitor for output holding time is reduced so that the static capacity can be decreased, and the conduction period of the input current from an AC power source can be broadened. Also, the peak value of such current can be suppressed. Thus, an excellent switching power supply device which exhibits improved power factor and improved efficiency can be obtained.

Claims: A switching power supply device comprising a DC power source and a transformer (3), said device further comprising: a first switching means (4, 5) connected in series with the primary winding (3a) of the transformer, this series circuit being connected to the terminals of the DC power source, a second switching means (11, 12), a capacitor (13) connected in series with the second switching means (11, 12), the series circuit including said capacitor (13) and said second switching means (11, 12) is connected across the primary winding (3a), a rectifying and smoothing circuit (7, 8), and a drive means (21) for driving alternately said first and second switching means (4, 5, 11, 12) with predetermined ON-OFF periods, said drive means (21) is formed to respond to the DC output voltage from said rectifying and smoothing circuit (7, 8), wherein: said transformer (3) further includes a bias winding (3b); said rectifying and smoothing circuit (7, 8) is formed for rectifying and smoothing a flyback voltage, which is generated in the secondary winding (3c) of the transformer and for outputting a resulting DC voltage; and said drive means (21) includes a first drive means (6) for driving said first switching means (4, 5) being connected to the bias winding (3b) of the transformer and a second drive means (9) for driving said second switching means (11, 12) in response to said DC output voltage. A switching power supply device according to claim 1, characterized in that at least one resonance capacitor (16, 17) is connected to the terminals of at least one of said first and second switching means (4, 5, 11, 12) and said primary and secondary windings (3a, 3c). A switching power supply device according to claim 1, characterized in that a) said DC power source comprises a rectifying circuit (19) for receiving and rectifying an AC voltage, said rectifying circuit (19) comprising at least one rectifying element, and an input capacitor means (20) for smoothing the output of said rectifying circuit (19), and b) said series circuit including said first switching means (4, 5) and said primary winding (3a) is connected to said input capacitor means (20). A switching power supply device according to claim 1, characterized in that at least one of said first and second switching means (4, 5, 11, 12) comprises a switching element (4, 11) and a diode means (5, 12) which are connected in parallel.

Inventors: ISHII TAKUYA, MAEOKA TATSUO, ISHII, TAKUYA, MAEOKA, TATSUO
Applicants: MATSUSHITA ELECTRIC IND CO LTD, MATSUSHITA ELECTRIC INDUSTRIAL CO., LTD.
Application year: 1990
Main classification: H02M3
IPC codes: H02M3