AC motor control

Abstract: An AC motor control circuit particularly suitable for motors having permanent magnet rotors, and stators which are supplied with AC current from an inverter. A current detection (33) detects the DC current input to the inverter (29) and produces an output signal which is supplied to a power factor calculation circuit (36). The output from this circuit is supplied to a voltage and frequency calculation circuit (38) whose output is in turn supplied to a drive circuit (38) which produces pulse-width-modulated drive signals for the transistors of the inverter (29).

Description: This invention relates to a control device for an AC motor using an inverter circuit. Control devices for AC motors may be of very many different types, depending on the type of the AC motor to be controlled. This invention is concerned with a device for controlling induction motors and brushless motors equipped with a rotor of the permanent magnet type. First of all, a device for controlling a brushless motor will be described. To control brushless motors, instead of employing a position detection element such as a Hall element to detect the relative position of the stator winding and permanent magnet type rotor, a system has come to be employed wherein this position is detected by using the terminal voltages, which contain the induced voltages generated in the stator winding. A prior art example of this is shown in Fig. 5 1 is a DC current source. 2 is an inverter circuit for passing current to stator windings 3U, 3V and 3W of brushless motor 3. 4, 5 and 6 are filter curciut that shift by 90 degress the phase of terminal voltages UV, VV, and WV containing the induced voltages generated in stator windings 3U, 3V and 3W. 7 is a detection circuit whereby the neutral point voltage NV is obtained from the output signals of these filter circuits 4 to 6. 8, 9 and 10 are comparators that respectively compare the output signals of filter circuits 4 to 6, which constitute first order delay elements, and neutral point voltage NV. 11 is a control circuit. Fig. 6 is a time chart showing the operation of the prior art example. We shall now consider the U phase with reference to this time chart. During commutation of the inverter circuit 2 the terminal voltage UV (see Fig. 6 (a)) generated in stator winding 3U contains a voltage spike that is produced by conduction of the opposite arm return flow diode. In order to eliminate the effect of this voltage spike component, the terminal voltage UV is shifted in phase by 90 degrees by means of filter circuit 4, producing the phase-shifted voltage DUV as shown in Fig. 6 (b). After this, this phase-shifted voltage DUV and the neutral point voltage NV shown in Fig. 6 (b) are compared by comparator 8, to obtain a phase detection signal PSU as shown in Fig. 6C. The situation is the same for the other phases V and W, position detection signals PSV and PSW being obtained as shown in Fig. 6(d) and (e) from comparators 9 and 10, based on terminal voltages VV and VW. These position detection signals PSU, PSV and PSW are signals that are 120 degrees different in phase for 180 degrees conduction. By applying these to control circuit 11, this control circuit 11 is made to produce six commutation signals, which are applied to the bases of the transistors that constitute the switching elements of inverter circuit 2. However, in the device described above for controlling a brushless motor, since filter circuits 4 to 6 having a 90 degrees lagging phase characteristic are provided in order to remove the voltage spike component contained in terminal voltages UV, VV and VW, the time constants of filter circuits 4 to 6 are large. This gives rise to the problem that it is not possible to track rapid acceleration or deceleration. A further problem is that phase detection in the low speed region is difficult. Furthermore, the size of the voltage spike component contained in terminal voltages UV, VV and WV varies depending on the magnitude of the current, i.e., the load of stator windings 3U, 3V and 3W, so if the load fluctuation is large, a phase error is produced in the signal waveform of filter circuits 4 to 6 et seq, causing a stability problem. Next, a device for controlling a three-phase induction motor will be described. This type of conventional device is disclosed Japanese Laid Open Patent Publication No. sho. 62-100192, and is illustrated in Fig. 7 A voltage type PWM inverter 12 consists of a rectifier circuit 14 that rectifies the three-phase AC voltage of three-phase AC power source 13, a smoothing capacitor 15 that smooths this rectified voltage, and a main inverter circuit 16 to which this smoothed DC voltage is applied. The AC output voltage from main inverter circuit 16 is then applied to three-phase induction motor 17. The current Idc flowing in the DC bus of inverter 12 is detected by current detector 18 and supplied as detection current I to low pass filter (hereinbelow termed "LPF")19. LPF 19 extracts the fundamental wave of the detected current I and outputs it as frequency correction value DELTA f. Subtractor 20 subtracts this frequency correction value DELTA f from the frequency command value f* to give a reference frequency value f (= f* - DELTA f) , which is applied to a pulse width modulation (PWM) control circuit 21. This pulse width modulation control circuit 21 is supplied with a voltage command value V* obtained by converting the frequency command value f* by means of a frequency-voltage (f - V) conversion circuit 22. As a result, pulse width modulation control circuit 21 performs pulse width modulation control by applying a base signal to the power transistors of main inverter circuit 16, based on reference frequency value f and voltage command value V*. Thus, since, in a voltage type PWM inverter, the DC voltage is fixed, the mean value of the current flowing in the DC bus of inverter 12 is proportional to the power supplied to three-phase induction motor 17. In this case, assuming that the rotational speed of three phase induction motor 17 is sufficiently high, and that the rate of change of the speed of rotation is very small, torque fluctuation is proportional to power fluctuation, so, by controlling the mean current of the DC bus of inverter 12, torque control of three-phase induction motor 17 can be performed, and production of vibration can be prevented. However, although the above control device is effective when the speed of revolution of the three phase induction motor 17 is sufficiently high, and the ratio of variation of speed of revolution is very small, in low speed operation, when the speed of revolution is low, the amount of power change for a given change of torque falls off. Even if this is therefore compensated by dividing by the speed of rotation or by the inferred value of the speed of rotation so as to remove dependence on the speed of operation, the lowered S/N ratio means that sufficient accuracy is not obtained. Stable driving of the inverter during low speed operation is therefore difficult. This invention was made after consideration of the above circumstances. Its first object is to provide a control device for an AC motor providing good response characteristics and improved stability during sudden accelerations or decelerations and load fluctuations, by controlling the AC motor in accordance with the amount of change of current. Its second object is to provide a control device for an AC motor whereby the AC motor can be driven in a stable manner even during low speed operation. According to the present invention there is provided a control device for an AC motor comprising inverter means including switching elements whereby DC voltage from DC power source is converted to AC voltage and supplied to said AC motor, characterised by; current detection means for detecting a current supplied from the inverter means to the AC motor; power calculation means for calculating the powers passed to said AC motor in two phases from a phase and a voltage command signal and the detection signal of the current detection means; and control means for supplying the phase and the voltage command signals to the power calculation means, to means for generating drive signals for the inverter means, such that the difference between the powers is zero or a predetermined value. Some embodiments of the invention will now be described by way of example with reference to the accompanying drawings in which: Fig. 1 is a block diagram showing the overall electrical layout of a first embodiment; Fig. 2 is a vector diagram given in explanation of the principles of operation of the first embodiment; Fig. 3 is a block diagram showing an example layout of a control circuit of a second embodiment of the invention; Fig. 4 is a block diagram showing an example layout of the control circuit of a third embodiment of the invention; Fig. 5 is an electrical layout diagram of a first example of a prior art control circuit; Fig. 6 is a signal waveform diagram of the various parts of the first prior art example; Fig. 7 is a block diagram showing a second example of prior art. First of all, the overall layout will be described with reference to Fig. 1. An inverter circuit 100 consists of a three-phase bridge connection of six switching elements constituted by transistors. This is supplied with the DC voltage of DC power source 101 at its input terminal. A three-phase four-pole brushless motor 102, is equipped with a stator 103 provided with U, V and W phase stator coils 103U, 103V and 103W, and with a permanent magnet type rotor 104. Stator coils 103U, 103V and 103W are connected in a star connection, which is supplied with AC output voltage from the output terminals of inverter circuit 100 through output leads 105, 106 and 107. 108 and 109 are current detectors consisting of Hall current transformers. These are respectively provided in the output leads 105 and 106 for the U phase and V phase. 110 is an amplifier circuit. Together with current detectors 108 and 109, this constitutes a current detection means 111. Current signals Iu and Iv detected by current detectors 108 and 109 are supplied to the input terminals of a power calculating means constituted by power calculation circuit 112 through amplifier circuit 110. As will be described, this power calculation circuit 112 uses current signals Iu and Iv to calculate power signals W1 and W2. These power signals W1 and W2 are output from its output terminal and supplied to the input terminal of a control moans constituted by a control circuit 113. This control circuit 113, as will also be described, uses power signals W1 and W2 and a speed command value omega* that is supplied from a setting device 114 to calculate voltage signal V and phase signal . These are in turn used to produce six PWM-controlled drive signals DS that are output from its output terminals and supplied to the bases of the six transistors of inverter circuit 100, causing the rotor 104 of brushless motor 102 to rotate with a speed of rotation designated by speed command value omege*. The principles of operation of this embodiment will now be explained with reference to Fig. 2 Fig. 2 is a vector diagram showing the relationship of the phase voltage, phase current and line voltage in each phase. Eu, Ev and Ew are the U, V and W phase voltages, Iu, Iv and Iw are the U, V and W phase currents, and  is the phase difference between the respective phase voltages Eu, Ev and Ew and the respective phase currents Iu, Iv and Iw. Euw and Evw are the line voltages referred to the W phase voltage Ew. Line voltage Euw has a phase difference (-π/6) with respect to the phase voltage Ev, and the line voltage Evw has a phase difference (+ π/6) with respect to the phase voltage Ev. When, from the above values, the power W1 and W2 are calculated by finding the product of the line voltage Euw and phase current Iu and the product of the line voltage Evw and phase current Iv, we obtain: W1 = Evw X Iv X cos ( -π/6) W2 = Evw X Iv X cos ( +π/6) From the vector diagram shown in Fig. 19, converting the above equations(24) and (25), we have: W1 = 3 X Ev X Iv X cos ( - π/6) W2 = 3 X Ev X Iv X cos ( + π/6) In this case, in the three-phase equilibrium condition, Eu = Ev = Ew and Iu = Iv = Iw, so, putting Eu = Ev = E and Iu = Iv = I, equations (26) and (27) above become: W1 = 3 X E X I X cos ( - π/6) W2 = 3 X E X I X cos ( + π/6) Finding the condition that power W1 and W2 are equal as between the two phases, we have: cos ( - π/6) = cos ( + π/6) = 0, Consequently, as described above, the power factor can be controlled by controlling the passage of current to stator coils 103U, 103V and 103W such that the power W1 and W2 of the two phases are equal, and brushless motor 102 can thus be driven in the samo way as conventionally. The detailed construction of cotrol circuit 113 for implementing the above operating principles will now be described with reference to Fig. 3 Speed command value omega* is applied to the respective input terminals of integrating element 116 and V/F control, element 117 through speed adjustment time element 115. Integrating element 116 outputs a phase signal from its output terminal and supplies this to the input terminal of drive control element 118. V/F control element 117 outputs a voltage command signal V from its output terminal and supplies this to one positive (+) input terminal of an adder 119. Adder 119 outputs a voltage signal V from its output terminal, as will be described. This voltage signal V is supplied to the input terminal of drive control element 118, and is also supplied to the input terminal of line voltage calculating element 120. Line voltage calculating element 120 receives at its other input terminal a phase signal  and uses this phase signal  and voltage signal V to calculate line voltages Euw and Evw, which it supplies as signals to the input terminals of power calculation circuit 112. Power signals W1 and W2 from power calculation circuit 112 are respectively supplied to the positive (+) input terminal and negative (-) input terminal of subtractor 121, and the output signal from the output terminal of the subtractor 121 is supplied through a PI (proportional/integral) control element 122 to the other positive (+) input terminal of adder 119. The operation of this embodiment will now be described. In control circuit 113, when speed command value omega* is applied to speed adjustment time element 115, the output signal omega* of speed adjustment time element 115 rises for a speed adjustment time that can be specified at will by speed command value omega*. This is then integrated by integrating element 116 and output as phase signal . It is also compared with a V/F pattern, which can be chosen at will, by V/F control element 117, and output as voltage command signal V. The voltages to be applied to stator coils 113U and 113W can then be calculated from the phase signal  that is output from integrating element 116 and the voltage signal V that is output from adder 119 that adds a correction to voltage command signal V. Line voltage calculating element 120 calculates these as line voltages Euw and Evw and applies them as signals to power calculation circuit 112. Power calculation circuit 112 calculates powers W1 and W2 as shown by equations (24) and (25) from the phase current signals Iu and Iv that are supplied from current detection means 111 and these line voltages Euw and Evw. These powers W1 and W2 are then fed back to control circuit 113 as signals and supplied to subtractor 121. Subtractor 121 finds the difference between these power signals W1 and W2 and supplies this difference signal to PI control element 122. PI control element 122 performs PI control such that this difference signal is zero or a prescribed value, its output being supplied to adder 119. Adder 119 outputs a voltage signal v by adding the output signal from PI control element 122 to voltage command signal V. The phase signal  and voltage signal v which are thus obtained are supplied to drive control element 118. Drive control element 118 uses this phase signal  and voltage signal v to output drive signals DS which it supplies to the bases of the six transistors of inverter circuit 100, thereby causing the rotor 104 of brushless motor 102 to rotate at a speed corresponding to the speed command value omega*. Thus, with this embodiment, the currents flowing in the U phase and V phase stator coils 103U and 103V are detected by current detection means 111, and these detected current signals Iu and Iv are used by the power calculation circuit 112 to calculate powers W1 and W2. Control circuit 113 then obtains a voltage signal v and phase signal  by PI control such that the difference of the powers W1 and W2 obtained as a result of this calculation is zero or a prescribed value. This voltage signal v and phase signal  are then in turn used by inverter circuit 100 to control operation of the brushless motor 102. As a result, in contrast to the coventional arrangement using the terminal voltage of the stator coils, it is unnecessary to use a first order delay element constituted by a filter circuit. This results in good response to abrupt accelerations or decelerations and load fluctuations, and enables stability to be raised. Also, this embodiment has excellent response to external disturbances, detection is possible in the low speed region, and the range of control can be increased. Fig. 4 shows another embodiment of this invention. Parts which are the same as in Fig. 3 are given the same refernce numerals. The difference with respect to Fig. 3 lies in that control circuit 113 is additionally provided with an adder 123 and P control elemet (or it could be a PI control element) 124. A phase signal  from integrating element 116 is applied to one of the positive (+) input terminals of adder 123, and a difference signal from adder 121 is applied to the other positive (+) input terminal of adder 123 through P control element 124, so that a phase signal  is output from the output terminal of adder 123. With this embodiment, even more stable feedback It should be noted that, although in the above embodiments 8 and 9 the drive devices were depicted by blocks of respective functions, for example power calculating circuit 112 and control circuit 113 could alternatively be implemented in a microcomputer. Although the invention has been described above with reference to specific embodiments, this invention is not restricted to the embodiments described above and illustrated in the drawings. For example, it is not restricted to three-phase motors but could be applied to other A.C. motors such as multi-phase brushless motors or induction motors in general, and could of course be put into practice in various modified ways within the scope of the appended claims.

Claims: A control device for an AC motor (102) comprising inverter means (100) including switching elements whereby DC voltage from DC power source (101) is converted to AC voltage and supplied to said AC motor (102), characterised by; current detection means (110) for detecting a current supplied from the inverter means (100) to the AC motor (102); power calculation means (112) for calculating the powers passed to said AC motor (102) in two phases (W1, W2) from a phase () and a voltage (V) command signal and the detection signal of the current detection means (110); and control means (113) for supplying the phase and the voltage command signals to the power calculation means, (112, 120) to means (118) for generating drive signals for the inverter means (100), such that the difference between the powers (W1, W2) is zero or a predetermined value.

Inventors: MOCHIKAWA HIROSHI, NAKAMURA YOSHINOBU, MOCHIKAWA, HIROSHI, NAKAMURA, YOSHINOBU, MOCHIKAWA, HIROSHI, C/O INTELL.PTY.DIV., NAKAMURA, YOSHINOBU, C/O INTELL.PTY. DIV.
Applicants: TOSHIBA KK, KABUSHIKI KAISHA TOSHIBA
Application year: 1990
Main classification: H02P7
IPC codes: H02P23, H02P27