Analog filter

Abstract: An analog filter for realizing the secondary transfer function by using an amplifier having a feedback loop. The filter has a first integrating function block (13) which integrates an input signal, second and third integrating function blocks (12, 14) which integrate an output signal, a first signal combining circuit (1) which combines signals from the first and second integrating function blocks (13, 12) with the input signal, a fourth integrating function block (11) which integrates signals from the first signal combining circuit (13) and a second signal combining circuit (2) which combines signals from the third and fourth integrating function blocks (14, 11) with the input signal and derives an output signal.

Description: ANALOG FILTER The present invention relates generally to an analog filter, and more particularly, to an analog filter which carries out a filter operation on analog signals by use of a feedback. An active system, such as a biquad system is known in the art for constituting analog filters. An analog filter made according to the biquad system is suitabe for implementing in an integrated circuit (referred to as IC hereafter). The biquad filter is used to get a desired filter characteristic by forming a circuit which realizes the secondary transfer function without using inductors but using an amplifier having a feedback loop. Such an analog filter, e.g., a biquad filter is widely used in color television fields, for instance, a SECAM system color television system, in a form of a bell filter. The bell filter has a transfer function as expressed as a following equation: G S = H · S 2 + ω 0 Q N · S + ω 0 2 S 2 + ω 0 Q P · S + ω 0 2 where, ω0 represents a natural angle frequency (e.g., 4.286 MHz); H represents a gain coefficient (actual number); S represents a complex number; Qp represents a pole selectivity or a selectivity assigned for the denominator of the equation (1); and Qn represents a zero selectivity or a selectivity assigned for the numerator of the equation (1). When assuming the gain G at the natural angle frequency of 0 dB, the gain coefficient H becomes Qn/Qp. To embody the transfer function various biquad filters have been considered. For instance, Y. Yamamoto et al., IEEE Transaction on Consumer Electronics, Vol. 3, August 1988, pp. 443-451 show such a biquad filter (see Figs. 3 and 5) and a video processor for color TV (see Fig. 2) using such a filter. However, such a conventional biquad analog filter has problems as follows. That is, the filter can not totally complete a differential operation. It is influenced by a parasitic capacitance when it is implemented on an IC device. Further, it requires an impedance transformer for driving capacitors in integration circuits constituting the filter. The present invention therefor seeks to to provide an analog filter which can reduce the above problems in the conventional filters. According to a first aspect of the present invention, there is provided an analog filter for realising a second order transfer function comprising: an input (X); an output (Y); first and second transconductance amplifiers (34, 32) having their respective inputs coupled to the output (Y) of the analog filter, the output of said first transconductance amplifier (34) being connected to ground by means of a first capacitor (C4), and the output of said second transconductance amplifier (32) being connected to ground by means of a second capacitor (C2); and a third transconductance amplifier (31) whose input is coupled to the common connection between the output of the second transconductance amplifier (32) and the second capacitor (C2) and whose output is coupled to the common connection between the output of the first transconductance amplifier and the first capacitor (C4); wherein the output of the third transconductance amplifier (31) is coupled to the output (Y), and the input (X) is coupled to the output (Y). According to a second aspect of the present invention, there is provided an analog filter for realising a second order transfer function comprising: a signal input (X); a signal output (Y); first and second transconductance amplifiers (34, 32) having their respective inputs coupled to the output (Y) of the analog filter, the output of the first transconductance amplifier (34) being connected to ground by means of a first capacitor (C4), and the output of the second transconductance amplifier (32) being connected to ground by means of a second capacitor (C2); a third transconductance amplifier (31) whose input is coupled to the common connection between the output of the second transconductance amplifier (32), and the second capacitor (C2), and whose output is coupled to the common connection between the output of the first transconductance amplifier (34) and the first capacitor (C4); wherein the output of the third transconductance amplifier (31) is coupled to the output (Y), and the input (X) is coupled to the input of the third transconductance amplifier (31). For a better understanding of the present invention and many of the attendant advantages thereof reference will now be made by way of example to the accompanying drawings, wherein: Figure 1 is a functional illustration showing an analog filter of related art; Figure 2 is a circuit diagram showing the filter of Figure 1; Figure 3 is a functional illustration showing an analog filter according to the present invention; Figure 4 is a circuit diagram showing the analog filter of Figure 3; FIGURE 5 is another circuit diagram showing the analog circuit of FIGURE 3 with a differential arrangement; FIGURE 6 is a functional illustration showing a low-pass filter as a first embodiment of the analog filter according to the present invention; FIGURE 7 is a circuit diagram showing the low-pass filter of FIGURE 6; FIGURE 8 is a functional illustration showing a high-pass filter as a second embodiment of the analog filter according to the present invention; FIGURE 9 is a circuit diagram showing the high-pass filter of FIGURE 8; FIGURE 10 is a functional illustration showing a notch filter as a third embodiment of the analog filter according to the present invention; FIGURE 11 is a circuit diagram showing the notch filter of FIGURE 10; FIGURE 12 is a functional illustration showing a band-pass filter as a fourth embodiment of the analog filter according to the present invention; FIGURE 13 is a circuit diagram showing the band-pass filter of FIGURE 12; FIGURE 14 is a functional illustration showing an all-pass filter as a fifth embodiment of the analog filter according to the present invention; and FIGURE 15 is a circuit diagram showing the all-pass filter of FIGURE 14. The present invention will be described in detail with reference to FIGURES 1 through 15. Throughout the drawings, like or equivalent reference numerals or letters will be used to designate like or equivalent elements for simplicity of explanation. Prior to the detailed descriptions of the present invention, an analog filter of related art will be described in brief in reference to FIGURES 1 and 2. FIGURES 1 and 2 show a fuctional arrangement and a circuit diagram of an analog filter, i.e., a biquad filter which I recently developed, but which was not publicly known at the time of this application. In FIGURE 1, Tin is an input terminal for receiving an input signal X, while Tout is an output terminal for outputting an output signal Y. This filter has two integrating function blocks 61 and 62, three signal combining function blocks 51, 52 and 53, three feed-forward passes L1, L2 and L3, which feed the input signal X with a prescribed gain, and three feedback passes L4, L5 and L6, which feed back the output signal Y with a prescribed gain. The feed-forward pass L1 includes a coefficient function block 71 with a coefficient of H. The feed-forward pass L2 includes a coefficient function block 72 with a coefficient of H/Qn. The feed-forward pass L3 has a coefficient of 1. The feedback pass L5 includes a coefficient function block 73 with a coefficient of [1 - 1/Qp]. The feedback passes L4 and L6 have a coefficient of 1, repectively. The analog biquad filter of FIGURE 1 can well represent the function of the above-mentioned Equation (1). The functional arrangement of the analog biquad filter, as shown in FIGURE 1, can be realized by a practical circuit as shown in FIGURE 2. In FIGURE 2, differential type transconductance amplifiers (hereinafter referred to as Gm amplifiers) 81 and 82 correspond to the intgrating function blocks 61 and 62, respectively. The input signal X on the input terimal Tin is applied to the non-inversed input terminal (+) of the front stage Gm amplifier 81 via an amplifier 83 which performs an impedance conversion as well as the function of the coefficient function block 71. An output of the Gm amplifier 81 is then transmitted to the non-inversed input terminal of the rear stage Gm amplifier 82. An output of the rear stage Gm amplifier 82 is lead to the output terminal Tout through an impedance conversion amplifier 84 and results to the output signal Y. A first capacitor 91 corresponding to the pass L2 of FIGURE 1 is coupled between the input terminal Tin and a first connection node of the output terminal of the front stage Gm amplifier 82 and the non-inversed input (+) of the rear stage Gm amplifier 82. A second capacitor 92 corresponding to the pass L2 of FIGURE 1 is coupled between the output terminal Tout and the first connection node. A third capacitor 93 corresponding to the pass L3 of FIGURE 1 is coupled between the output terminal of the amplifier 83 and a second connection node of the output terminal of the rear stage Gm amplifier 82 and the impedance conversion amplifier 84. The outupt signal Y is fed back to the inversed input terminals (-) of the Gm amplifiers 81 and 82, respectively. The capacitors 91, 92, 93 and the impedance conversion amplifier 84 define the gains of the passes L1 through L6 in FIGURE 1. When assuming that the transconductance of the front stage Gm amplifier 81 is Gm1, the transconductance of the rear stage Gm amplfier 82 is Gm2, and the capacitances of the capacitors 91, 92 and 93 are C1, C2 and C3, respectively, then the transfer function T(S) of the analog filter of FIGURE 2 is expressed by the following equation: T (S) = H· S 2 + C 1 ·g m2 H· (C 1 + C 2 ) C 3 S + g m1 ·g m2 (C 1 + C 2 )C 3 S 2 + C 1 ·g m2 (C 1 + C 2 ) ·C 3 S + g m1 · g m2 (C 1 + C 2 ) C 3 The parameters ω0, Qp and Qn in Equation (1) are expressed as follows, from the comparison of Equations (1) and (2): ω 0 = g m1 · g m2 (C 1 + C 2 ) · C 3 Q P = (C 1 + C 2 ) · C 3 C 1 2 · g m1 g m2 Q N = H (C 1 + C 2 ) · C 3 C 1 2 · g m1 g m2 When assuming that C1 + C2 = C3 = C0, Gm1 = Gm2 = Gm0, C 0 = g m0 ω 0 C 1 = 1 Q P C 0 C 2 = (1 - 1 Q P ) C 0 C 3 = C 0 H = Q N Q P As seen from the above, Equations (4a) to (4d) have the ratio forms which are suitable for ICs, the parameters ω0, Qp and Qn in Equations (3a), (3b) and (3c) are determined based on the capacitance ratios. Thus, the analog filter of FIGURES 1 and 2 is suitable for implementing on ICs and presents extremely excellent characteristics. However, the inventor found from experiments on practical filters made on an IC substrate that the analog filter of FIGURES 1 and 2 still had deteriorations as described below. A first problem is a problem caused by parasitic capacitances. There are such parasitic capacitances as collector substrate capacitances of transistors constituting the Gm amplifiers, a collector base capacitance, a wiring capacitance, etc. These parasitic capacitances cause a large influence when they are formed on high impedance regions. In FIGURE 2, parasitic capacitances CP2 and CP3 formed at positions shown by a dotted line cause the problems. The parasitic capacitance Cp2 is formed between the first connection node and a ground circuit. The parasitic capacitance Cp3 is formed between the second connection node and the ground circuit. When such parasitic capacitances CP2 and CP3 are formed, the transfer function Tp(S) is expressed by the following equation; T P (S) = C 3 C 3 + C p3 H× S 2 + C 1 · g m2 (C 1 + C 2 + C p2 ) · C 3 S + g m1 · g m2 (C 1 + C 2 + C p2 ) · C 3 S 2 + (C 1 + C p2 ) · g m2 (C 1 + C 2 + C p2 )(C 3 + C p3 ) S + g m1 · g m2 (C 1 + C 2 + C p2 )(C 3 + C p3 ) When Equation (5) is compared with Equation (1), the parameters become as follows; Q P = (C 1 + C 2 + C p2 )(C 3 + C p3 ) (C 1 + C p2 ) 2 · g m1 g m2 Q N = (C 1 + C 2 + C p2 ) C 1 2 · g m1 g m2 ω P = g m1 · g m2 (C 1 + C 2 + C p2 )(C 3 + C p3 ) ω N = g m1 · g m2 (C 1 + C 2 + C p2 ) · C 3 As seen from the above equations, the parasitic capacitance Cp3 relates to the equation (6c) for the pole natural angle frequency ωP, but do not relate to the equation (6d) for the zero point natural angle frequency ωN. Therefore, it is impossible to make the polar and the zero point natural frequencies ωP and ωN agree with the natural angle frequency ω0. Accordingly, when a biquad filter is constituted in such structure as shown in FIGURE 2, a deviation of the characteristic of the filter is unavoidable. A second problem is that although the Gm amplifiers 81 and 82 have a differential circuit configuration, two input signals applied to the non-inversed input terminal and the inversed input terminal of the respective Gm amplifiers 81 and 82 are not symmetrical with each other. When two asymmetrical input signals are input to the differential amplifier circuit such as the Gm amplifiers 81 and 82, its frequency characteristic becomes deteriorated. In the bell filter its pole and zero selectivities Qp and Qn are defined to have values of 16 and (1.6) 1/2 . That is, Qp = 16 and Qn = (1.6) 1/2 . Therefore, if the Gm amplifiers 81 and 82 do not operate differentially, out of phase as an integration circuit will cause a large influence and the filter characteristic will deviate. A third problem is that some capacitors are driven by voltage signals because the amplifier 83 with the gain of H and the amplifer 84 with the gain of 1 are a type of voltage signal outputting circuit, i.e., the amplifiers 83 and 84 have a very low impedance. Thus, the Gm amplifiers 81 and 82 exhibit a low-pass filter characteristic. Therefore, phase lags occur based on the output impedances of the amplifiers 83, 84 and the capacitances C3, C2 so that a deviaion of the filter characteristic arises. Especially, in filters having a very large value of selectivity Q such as the bell filter, a phase lag in the amplifier 84 with the gain of 1 which is connected to the feedback passes L4, L5 and L6 largely affects on its filter characteristic. In the excessive case, an oscillation may occur in the filter. Referring now to FIGURES 3 and 4, a first embodiment of the analog filter according to the present invention will be described in detail. The analog filter of FIGURES 3 and 4 is accomplished by farther considerations for improving the analog filter of FIGURES 1 and 2. In FIGURE 3 the first embodiment of the analog filter is constituted in a form of the bell filter. Feed-forward passes L11 and L12 for feeding forward an input signal X to be filtered and the feedback passes L13 and L14 for feeding back an output signal Y of the analog filter are connected to a main pass connected between an input terminal Tin and an output terminal Tout of the analog filter for applying a second order function on the input signal X. In this main pass, a first integrating function block 13, a first signal combining function block 1, a fourth integrating function block 11 and a second signal combining function block 2 are connected in series. The first integrating function block 13 multiplies by H times the input signal X as well as integrates the input signal X. The first signal combining function block 1 combines the output from this first integrating function block 13 and signals from the passes L12 and L13 together, wherein the signal from the pass L12 is applied to the first signal combining function block 1 in the opposite phase to the phases of the other two signals. The fourth integrating function block 11 integrates the output from the first signal combining function block 1. The second signal combining function block 2 combines the output from the fourth integrating function block 11 and signals from the passes L11 and L14 together, wherein the signal from the pass L14 is applied to the second signal combining function block 2 in the opposite phase to the phases of the other two signals. The output from the second signal combining function block 2 becomes the output signal Y. The feed-forward pass L11 has a coefficient function block 22 which multiplies H times the input signal X. The feed-forward pass L12 has a coefficient function block 21 which multiplies H(1/Qn) times, i.e., Qn times the input signal X. The feedback pass L13 has a second integrating function block 12 which integrates the output signal Y. The feedback pass L14 has a third integrating function block 14 which multiplies 1/Qp times the output signal Y as well as integrates the output signal Y. The above functional arrangement operates to accomplish the transfer function of Equation (1). Now, the operation of the FIGURE 3 for accomplishing Equation (1) will be explained. First, divide the denominator and numerater of Equation (1) by S², then a following equation (7) is obtained. G (S) = H · 1 + 1 Q N ( ω 0 S ) + ( ω 0 S ) 2 1 + 1 Q P ( ω 0 S ) + ( ω 0 S ) 2 = Y X Change Equation (7) as to its denominator, then a following equation (8) is obtained. {1 + 1 Q P ( ω 0 S ) + ( ω 0 S ) 2 } Y = H {1 + 1 Q N ( ω 0 S ) + ( ω 0 S ) 2 } X Move the third term in the left side of Equation (8) to the right side and rearrange it, then a following equation (9) is obtained. {1+ 1 Q P ( ω 0 S )}Y=H{1+ 1 Q N ( ω 0 S ) + ( ω 0 S ) 2 }X-( ω 0 S ) 2 Y =HX+ ( ω 0 S ) {H 1 Q N X+H( ω 0 S ) X-( ω 0 S ) Y} Move the second term in the left side of Equation (9) to the right side, then a following equation (10) is obtained. Y= HX+ ( ω 0 S ) {H 1 Q N X+H( ω 0 S )X-( ω 0 S )Y} - 1 Q P ( ω 0 S ) Y According to the above Equation (10), for instance, addition and subtraction arithmetic in parentheses, except the term of the input signal X, is calculated based on only current signals output from the integrating function blocks. Thus, capacitance elements typically used in the integrating function blocks are are driven by current signals only. Accordingly, the embodiment of FIGURE 3 requires no impedance conversion circuit so that phase lags are avoided. Furthermore, the arithmetic calculation for outputs from the integrating function blocks can be realized by simple line connections. Thus, the embodiment is not troubled by deviations of frequency characteristic caused by insufficient differential operation of differential circuits. Further, as an amplifier in each integrating function block operates on one input and one output, it is possible to improve the frequency characteristic by constituting a full differential circuit. FIGURE 4 shows a practical circuit for realizing the bell filter of FIGURE 3. The bell filter of FIGURE 4 is implemented on ICs by Gm amplifiers and capacitors. In FIGURE 4, the first Gm amplifier 33 corresponds to the first integrating function block 13 of FIGURE 3. The second Gm amplfier 32 corresponds to the second integrating function block 12. The third Gm amplfier 34 corresponds to the third integrating function block 14. The fourth Gm amplfier 31 corresponds to the fourth integrating function block 11. The transconductances of the first and fourth Gm amplifiers 33 and 31 are expressed in the positive expression, while the transconductances of the second and third Gm amplifiers 32 and 34 are expressed in a negative expression. Capacitors 41 through 44 constitute integrating capacitances to the Gm amplifiers 31 through 34. The capacitor 41 is connected between the input terminal Tin and a first connection node of the first and fourth Gm amplifiers 33 and 31. The capacitor 42 is connected between the first connection node and the ground circuit. The capacitor 43 is connected between the input terminal Tin and a second connection node of the fourth and second Gm amplifier 31 and 32. The capacitor 44 is connected between the second connection node end the ground circuit. When the transconductances of the above Gm amplifiers 31 through 34 are assumed to be Gm1, Gm2, Gm3 and Gm4, and the capacitances of the capacitors 41 through 44 to be C1, C2, C3 and C4, the transfer function T(S) of the bell filter of FIGURE 4 is expressed as follows: T(S) = C 3 C 3 + C 4 · S 2 + C 1 · g m1 (C 1 + C 2 )C 3 S + g m1 · g m3 (C 1 + C 2 )C 3 S 2 + g m4 C 3 + C 4 S + g m1 · g m2 (C 1 + C 2 )(C 3 + C 4 ) When Equation (11) is compared with Equation (1), the following parameters ωN, Qn, ωN and Qp are obtained: ω N = g m1 · g m2 (C 1 + C 2 ) C 3 ω P = g m1 · g m2 (C 1 + C 2 ) (C 3 + C 4 ) Q N = (C 1 + C 2 ) C 3 C 1 2 · g m3 g m1 Q P = (C 3 + C 4 ) (C 1 + C 2 ) · g m1 · g m2 g m4 2 From Equations (12), it is understood that the parameters of pole point Qp and zero point Qn are determined depending on the capacitance ratio and the transconductance ratio, like the bell filter of FIGURE 1, and thus that the construction of FIGURE 4 is suitable for incorporation into ICs and accurate values are obtainable. As a relation ωP = ωN = ω0 is obtained when assuming that Gm1 = Gm2 = Gm0 and C1 + C2 = C3 + C4 = C0, the following parameters are given by following equations (13a) through (13h). C 0 = g m0 ω 0 C 1 = 1 Q P C 0 C 2 = (1 - 1 Q P ) C 0 C 3 = Q N Q P C 0 C 4 = (1 - Q N Q P ) C 0 g m1 = g m2 = g m0 g m3 = Q N Q P g m0 g m4 = 1 Q P g m0 Now parasitic capacitances of FIGURE 4 will be discussed. In FIGURE 4, high impedance positions are the signal passes. As parasitic capacitances occur at these high impedance positions, a parasitic capacitance Cp2 is formed in parallel with the capacitor C2 while a parasitic capacitance Cp4 is formed in parallel with the capacitor C4 as shown in FIGURE 4. When such parasitic capacitances CP2 and CP4 are formed, the transfer function Tp(S) is expressed by the following equation; T P (S) = C 3 C 3 + C 4 + C p4 × S 2 + C 1 ·g m1 (C 1 + C 2 + C p2 )C 3 S + g m1 · g m3 (C 1 + C 2 + C p2 ) C 3 S 2 + g m4 C 3 + C 4 + C p4 S + g m1 · g m2 (C 1 + C 2 + C p2 )(C 3 + C 4 + C p4 ) Equation (14) is basically equal to Equation (11). That is, (C2 + Cp2) in Equation (14) corresponds to C2 in Equation (11) while (C4 + Cp2) in Equation (14) corresponds to C4 in Equation (11). This indicates that it is possible to remove the influence of the parasitic capacitances Cp2 and Cp4 by reducing the capacitances of the actual capacitics C2 and C4 by the parasitic capacitances Cp2 and Cp4. When viewed from a different point, Equation (12) indicates that the natural pole and zero frequencies ωP and ωN are settable independently in the bell filter of FIGURE 4 and the deviation of the transfer function caused by the parasitic capacitances Cp2 and Cp4 is adjusted to restore a prescribed characteristic. FIGURE 5 shows a modification of the bell filter of FIGURE 4. In FIGURE 5, the bell filter is constructed in the form of a full differential arrangement. According to such an arrangement, a differential circuit consisting of a pair of + Gm and - Gm amplifiers operates in a virtually grounded state. Therefore, its frequency characteristic is remarkably improved and the phase lag is greatly reduced. As a result, the bell filter of FIGURE 5 operates almost ideally. The deviation of the filter characteristic is also reduced. Further, each pair of the capacitors 42 and 44 which are coupled to the ground circuit can be regarded as virtually grounded. Therefore, each of them can be realized by one capacitor. In this case, it is sufficient if one capacitor having a capacitance less than half of that shown in FIGURE 5 is provided between the Gm amplfiers 31 and 32. This is further advantageous for implementing on ICs. The bell filter of FIGURES 3, 4 and 5 has no impedance conversion circuit, as described above. Thus, the bell filter can avoid an undesired ocillation, if the selectivity Q is especially high. Further, there is an avantage that the influence of the drive impedance of the bell filter is less even when the selectivity Q is high, because the capacitor C1 connected to the input terminal Tin is inversely proportional to Qp. It is further advantageous that the bell filter of FIGURES 3 and 4 realizes the double second order function which is a regular form of second order transfer functions. This suggests that all the second order transfer functions can be accomplished in many filters, not only the bell filter of FIGURES 3 and 4. A low-pass filter according to the present invention is shown in FIGURES 6 and 7. In FIGURES 6 and 7, respective elements which are similar to those in FIGURES 3 and 4 are assigned the same symbols as those shown in FIGURES 3 and 4. That is, the low-pass filter of FIGURES 6 and 7 has removed the coefficient function blocks 21 and 22 or the capacitors 41 and 43 from the bell filter of FIGURES 3 and 4. The transfer function TL(S) of the low-pass filter is expressed as follows: T L (S) = g m3 g m2 · g m1 · g m2 C 2 · C 4 S 2 + g m4 C 4 · S + g m1 · g m2 C 2 · C 4 A high-pass filter according to the present invention is shown in FIGURES 8 and 9. In FIGURES 8 and 9, respective elements which are also similar to those in FIGURES 3 and 4 are assigned the same symbols as those shown in FIGURES 3 and 4. That is, the high-pass filter of FIGURES 8 and 9 has removed the integrating function block 13 and the coefficient function block 21 or the front stage Gm amplifier 33 and the capacitor 41 from the bell filter of FIGURES 3 and 4. The transfer function TH(S) of the high-pass filter is expressed as follows: T H (S) = C 3 C 3 + C 4 · S 2 S 2 + g m4 C 3 + C 4 S + g m1 ·g m2 C 2 (C 3 + C 4 ) A notch filter according to the present invention is shown in FIGURES 10 and 11. In FIGURES 10 anhd 11, respective elements which are also similar to those in FIGURES 3 and 4 are assigned with the same symbols as those shown in FIGURES 3 and 4. That is, the notch filter of FIGURES 10 and 11 has removed the coefficient function block 21 or the capacitor 41 from the bell filter of FIGURES 3 and 4. The transfer function TN(S) of the notch filter is expressed as follows: T N (S) = C 3 C 3 + C 4 · S 2 + g m1 ·g m3 C 2 · C 3 S 2 + g m4 C 3 + C 4 S + g m1 · g m2 C 2 (C 3 + C 4 ) A band-pass filter according to the present invention is shown in FIGURES 12 and 13. In FIGURES 12 and 13, respective elements which are also similar to those in FIGURES 3 and 4 are assigned the same symbols as those shown in FIGURES 3 and 4. That is, the band-pass filter of FIGURES 12 and 13 has removed the integrating function block 13 and the coefficient function block 22 or the front stage Gm amplifier 33 and the capacitor 43 from the bell filter of FIGURES 3 and 4. The transfer function TB(S) of the band-pass filter is expressed as follows: T B (S) = C 1 ·g m1 (C 1 + C 2 ) ·g m4 · g m4 C 4 S S 2 + g m4 C 4 S + g m1 ·g m2 (C 1 + C 2 ) C 4 An all-pass filter according to the present invention is shown in FIGURES 14 and 15. In FIGURES 14 and 15, respective elements which are also similar to those in FIGURES 3 and 4 are assigned the same symbols as those shown in FIGURES 3 and 4. However the first signal combining function block 1 combines the output from the first integrating function block 13 and signals from the passes L12 and L13 together, wherein the signal from the integrating function block 13 is applied to the first signal combining function block 1 in the opposite phase to the phases of the other two signals. The fourth integrating function block 11 integrates the output from the first signal combining function block 1. Also the second signal combining function block 2 combines the output from the fourth integrating function block 11 and signals from the passes L11 and L14 together, wherein the signals from the pass L14 and the integrating function block 11 are applied to the second signal combining function block 2 in the opposite phase to the phase of the other signal from the coefficient function block 22. In the expression as shown in FIGURE 15, the transconductance of the second Gm amplifier 32 is expressed in the positive expression, while the transconductances of the first, third and fourth Gm amplifiers 33, 34 and 31 are expressed in the negative expression. The transfer function TA(S) of the all-pass filter is expressed as follows: T A (S) = C 3 C 3 + C 4 · S 2 - C 1 ·g m1 (C 1 + C 2 ) C 3 S + g m1 ·g m3 (C 1 + C 2 ) C 3 S 2 + g m4 C 3 + C 4 S + g m1 ·g m2 (C 1 + C 2 ) (C 3 + C 4 ) Thus, the transfer functions of the above embodiments are accomplished with all the second order transfer functions. Further, in each embodiment the passes to which the coefficient function blocks or capacitors are connected are changed depending on transconductances of the Gm amplfiers and in the case where a gain of the coefficient function block is 1, they are shown simply as passes on the drawings. As described above, the present invention can provide an extremely preferable analog filter. While there have been illustrated and described what are at present considered to be preferred embodiments of the present invention, it will be understood by those skilled in the art that various changes and modifications may be made, and equivalents may be substituted for elements thereof without departing from the true scope of the present invention. In addition, many modifications may be made to adapt a particular situation or material to the teaching of the present invention without departing from the central scope thereof. Therefore, it is intended that the present invention not be limited to the particular embodiment disclosed as the best mode contemplated for carrying out the present invention, but that the present invention include all embodiments falling within the scope of the appended claims.

Claims: An analog filter for realising a second order transfer function comprising: an input (X); an output (Y); first and second transconductance amplifiers (34, 32) having their respective inputs coupled to the output (Y) of the analog filter, the output of said first transconductance amplifier (34) being connected to ground by means of a first capacitor (C4), and the output of said second transconductance amplifier (32) being connected to ground by means of a second capacitor (C2); and a third transconductance amplifier (31) whose input is coupled to the common connection between the output of the second transconductance amplifier (32) and the second capacitor (C2) and whose output is coupled to the common connection between the output of the first transconductance amplifier and the first capacitor (C4); wherein the output of the third transconductance amplifier (31) is coupled to the output (Y), and the input (X) is coupled to the output (Y). An analog filter for realising a second order transfer function comprising: a signal input (X); a signal output (Y); first and second transconductance amplifiers (34, 32) having their respective inputs coupled to the output (Y) of the analog filter, the output of the first transconductance amplifier (34) being connected to ground by means of a first capacitor (C4), and the output of the second transconductance amplifier (32) being connected to ground by means of a second capacitor (C2); a third transconductance amplifier (31) whose input is coupled to the common connection between the output of the second transconductance amplifier (32), and the second capacitor (C2), and whose output is coupled to the common connection between the output of the first transconductance amplifier (34) and the first capacitor (C4); wherein the output of the third transconductance amplifier (31) is coupled to the output (Y), and the input (X) is coupled to the input of the third transconductance amplifier (31). An analog filter as claimed in claim 2 further comprising a fourth transconductance amplifier (33), the input of the fourth transconductance amplifier (33) being coupled to the input (X), and the output of the fourth transconductance amplifier being coupled to the input of the third transconductance amplifier (31). An analog filter as claimed in claim 3 further comprising a third capacitor (C3) coupled between the analog filter input (X) and the output (Y). An analog filter as claimed in one of claims 2-4, wherein the input (X) is coupled to the input of the third transconductance amplifier (31) by means of a fourth capacitor (C1). An analog filter as claimed in any preceding claim wherein the transconductance of the first and third transconductance amplifier (34, 32) is negative, and the transconductance of the second transconductance amplifier (31) is positive. An analog filter as claimed in claim 5 when dependent on claim 4 wherein the transconductance of the second transconductance amplifier is positive and the transconductance of the first, third and fourth transconductance amplifier is negative. An analog filter as claimed in claim 5 when dependent on claim 4 wherein the transconductance of the third and fourth transconductance amplifier is positive, and the transconductance of the first and second transconductance amplifier is negative. A differential analog filter comprising two analog filters as claimed in any one of claims 1-5, the first and second analog filters being identical apart from the transconductance of each transconductance amplifier in one filter being opposite in sign to that of the corresponding transconductance amplifier in the other filter, wherein the differential input signal is applied between the inputs of the first and second analog filters and the differential output signal is obtained from the outputs of the first and second analog filters. A differential analog filter as claimed in claim 9 wherein each pair of the first and second capacitors is replaced by a single capacitor connected between the analog filters.

Inventors: NAKAGAWARA CHIKASHI, NAKAGAWARA, CHIKASHI, NAKAGAWARA, CHIKASHI, C/O INTELLECTUAL PROP. DIV.
Applicants: TOSHIBA KK, KABUSHIKI KAISHA TOSHIBA
Application year: 1990
Main classification: H03H11
IPC codes: H03H11