Metrological apparatus and calibration method therefor

Abstract: A metrological apparatus for measuring form and texture of a surface employs an inductive transducer producing an AC gauge signal, and a tracking AC analogue to digital converter. In order to avoid errors in the output signal at a relatively high speed of traverse of the surface due to a non-flat frequency response of the converter, the output signal is processed by a digital filter to provide a combined frequency response which is flat over a desired range. In order to avoid errors in the output signal due to non-linearity of the transducer, a square or cubic correction is applied to the signal. A method calibrating the apparatus to obtain coefficients for the square and cube terms in the correction process is also described.

Description: This invention relates to metrological apparatus and particularly concerns such apparatus for measuring surface characteristics of a workpiece. Conventionally, such apparatus includes a stylus which traverses the surface of the workpiece and a transducer which produces a signal dependent upon the movement of the stylus in response to surface characteristics such as texture or form. Most presently available apparatus for measurement of surface characteristics are only capable of measuring surface texture, which requires high resolution, in a separate operation from the measurement of form, requiring a relatively long range of operation of the transducer. This is because most currently available apparatus arranged for sensing the surface of a workpiece have a relatively low ratio of range to resolution. An instrument which is capable of simultaneously measuring surface texture and form is commercially available but this employs an interferometric transducer which provides the required high ratio of range to resolution and such a transducer is expensive and, due to the need to accommodate various optical elements and a laser, the apparatus is somewhat bulky. Relatively low cost transducers are available, in particular, transducers which operate inductively. These transducers produce an analogue output signal which, in order to be useful in modern computer based instruments, must be converted to digital form. Current gauging circuitry, at reasonable cost, would not justify the use of current high accuracy analogue to digital converters, with greater than 10 bit resolution. This does not provide sufficient resolution over the required range to permit surface texture and form to be measured simultaneously. Accordingly, there is a need for a low cost surface sensor arrangement which includes a transducer connected directly to an analogue to digital converter which is capable of converting the output of the transducer to digital form with the required range and resolution, thus obviating the need for current gauging circuitry with its shortcomings. Further, it is desirable that the instrument should be capable of operation at various speeds of traverse of the surface sensor. Therefore, to meet this need, it is also necessary that the analogue to digital converter should be capable of accurately digitising an analogue input signal whose frequency may vary dependent upon the speed of traverse of the sensor over the workpiece surface and upon the workpiece surface characteristics. There is known from GB-A-2207511 an electronic gauge in which an output variable (in this case the frequency of a signal) is non-linearly related to displacement distance owing to an S-shape error. This is corrected by quadratic correction equations. The co-efficients of the quadratic equations are said to be determined experimentally, but no indication is given of how this might be done. The purpose of the quadratic equations appears to be to improve the accuracy of the transducer over its normal range, rather than to extend its range. The transducer shown in GB-A-2207511 has a core which moves in a straight line, as it is in line with the stylus. This avoids errors arising from curved motion of the core, which can arise in a transducer in which the stylus and the core rotate around a pivot, so that the correction of transducer errors is relatively simple. However, the bulk of the in-line arrangement of the core prevents the transducer from being used to measure surfaces with restricted access, such as the inner surface of a ring. According to the present invention there is provided a metrological apparatus as set out in claim 1, a calibration method as set out in claim 10, and a method of measuring as set out in claim 12. The remaining claims set out optional features. An embodiment of the invention is described, by way of example, with reference to the accompanying drawings in which: Fig. 1 is a diagram of a metrological apparatus in which the invention may be embodied; Fig. 2 is a simplified block diagram of electrical circuitry included in the apparatus of Fig. 1; Fig. 3 shows, in more detail, part of the circuitry of Fig. 2; Fig. 4 is a block diagram showing in detail part of the circuitry of Fig. 3; Fig. 5 is a plot illustrating the frequency response of the circuitry of Figs. 3 and 4; Fig. 6 illustrates a transfer function of the circuitry of Figure 4; Fig. 7 illustrates a transfer function of the circuitry of figure 3; Fig. 8 schematically shows the operation of a filter associated with the circuitry of figure 3; and Fig. 9 is a diagram illustrating the movement of a stylus included in the apparatus of figure 1, in order that the calibration of the apparatus may be understood. With reference to Figs. 1 and 2, a metrological instrument for measuring surface texture or form comprises a base 2 for supporting a workpiece 4 having a surface 6 whose texture is to be measured. A vertical column 8 secured to the base 2 supports a vertically movable carriage 10 on which a horizontal datum bar 12 is mounted. A further carriage 14 is mounted for horizontal movement along the bar 12 and carries a transducer 16 having a stylus arm 18 carrying a stylus 19. A motor 20 indicated diagrammatically in Fig. 1 is provided on the carriage 14 for effecting horizontal movement thereof along the datum bar 12 and a further motor 21 is provided on the carriage 10 for effecting vertical movement thereof up and down the column 8. In operation of the apparatus, the carriage 10 is arranged at a vertical position such that the stylus 19 engages the surface 6. Thereafter, the motor 20 is operated to cause the carriage 14 to move horizontally along the datum 12 thus causing the stylus 19 to traverse the surface 6 to be measured. A linear optical grating, shown schematically at 22 in figure 1, comprising a series of reflective and non-reflective bands, is fixed with respect to the datum 12 and light from a light source 24 is reflected from the grating 22 and detected by a pair of light sensors 26, 27 mounted on the carriage 14. Alternatively, the grating may be formed of a series of transmissive and non-transmissive bands, with the sensors 26, 27 arranged to detect light transmitted through the grating. A reference scale (not shown) may be provided associated with the sensors 26, 27. As the carriage moves along the datum bar 12, the sensors 26, 27 provide sinusoidal electrical signals on lines 28 and 30 respectively. The sensors 26 and 27 are positioned so that the signals on lines 28 and 30 are in quadrature. These signals are applied to an interpolator 32 which causes their frequencies to be quadrupled. This may be done in the manner described in EP-A-0 404 598. The interpolator 32 outputs the resulting signals on lines 34 and 36 and supplies them to a zero crossing detector 38 which outputs a pulse on line 39 in response to each zero crossing point in each of the signals on lines 34 and 36. In one commercially available sensor arrangement, if the grating 22 has a pitch of 8 microns, the signals from sensors 26, 27 have a period corresponding to 4 microns. A pulse then appears on lines 39 for each 0.25 microns of movement of the carriage 14 along datum bar 12. As the carriage 14 is moved along the datum bar 12 and the stylus 19 is thereby caused to traverse the workpiece surface 6, the stylus 19 moves substantially vertically in response to the features of surface texture and form of the surface 6. Transducer 16 is an inductive transducer and, in response to this substantially vertical movement, produces an output signal which is digitised by an analogue-to-digital converter 40 and supplied to a microprocessor 42 which in response to the pulses on lines 39, samples the digitised signal from transducer 16 at 0.25 micron intervals of movement of the carriage 14 along the datum bar 12 and stores the resulting samples in a memory 44. The apparatus includes a computer 46 which is programmed to effect a general overall control of the apparatus, including control of motors 20 and 21, and to process the signals stored in memory 44 to provide information as to the surface texture and form of the surface 6. A keyboard 45 is provided for inputting instructions to computer 46. Signals representing the surface characteristics are output by computer 46, via a low pass filter 48 which filters out unwanted high frequency noise, to a display device 49, such as a cathode ray tube, and a printer 50 which may be operable to produce a trace of the surface of the workpiece. The display device 49 and printer 50 may receive digital input signals. Alternatively, either or both of them may receive analogue input signals provided via an optional digital-to-analogue converter 47 as shown in Figure 2. With reference to Fig. 3, the transducer 16 comprises a magnetic core 52 which is attached to one end of the stylus arm 18 and is movable in either direction as shown by the arrow 54 in response to movement of a stylus 19 at the other end of the stylus arm 18, as the stylus 19 traverses the workpiece surface. Coils 56, 58 magnetically coupled to the core 52 are centre tapped at 60 and form two arms of a bridge circuit, the other two arms of which are constituted by equal resistors 62, 64. A reference oscillator 65 supplies reference oscillations Vsinwt to the bridge circuit and the arrangement is such that, when the core 52 is at a centre position, the impedence of the coils 56 and 58 is equal and the voltage difference between the centre tap 60 and a further centre tap 66 between resistors 62 and 64 is zero. If the core 52 is displaced in one direction, an in-phase sinusoidal voltage kVsinwt appears between taps 60 and 66 having an amplitude kV (k being between 0 and 1) dependent upon the distance through which the core is moved. If the core 52 is moved in the opposite direction from the centre position, an out-of-phase voltage kVsinwt (k being between 0 and -1) of opposite phase appears between taps 60 and 66 with an amplitude -kV dependent upon the distance through which the core 52 has been moved. Taps 60 and 66 are connected to provide a gauge signal kVsinwt to input terminals 70, 72 of analogue-to-digital converter 40 which also receives a reference signal Vsinwt from the oscillator 65 on further input terminals 74, 76. The circuit 40 outputs a digital signal  representing the amplitude of a voltage between taps 60 and 66 for sampling by microprocessor 42. The analogue to digital converter 40 is operable to provide a high resolution output signal, preferably to 15 bits and is constructed as shown in Fig. 4. Referring to figure 4, the digital value of the output signal is held in an up-down counter 100. The gauge signal kVsinwt and the reference signal Vsinwt are input to buffer amplifiers 102, 104 respectively, and the buffered values are fed to a high accuracy multiplier 106. The multiplier 106 also receives the digital output value  from the counter 100 and is operable to multiply the gauge and reference signals by cos  and sin , respectively, to produce outputs on lines 108, 110 of kVsinwt.cos and V sinwt.sin, respectively. These signals on lines 108, 110 are then subtracted at an error amplifier 112 to produce an error signal Ve = Vsinwt (k cos - sin), which can be represented as Ve = V 1+k 2 sinwt.sin (-) where sin = k/ 1+k 2 . The error signal Ve is fed to a phase sensitive detector 114, where it is multiplied by the buffered reference signal Vsinwt to produce a signal Vpsd given by Vpsd = V 2 1+k 2 .sin 2 wt.sin (-) which can be represented as Vpsd = V 2 1+k 2 .sin(-).(1-cos2wt)/2 This signal is then filtered by a filter 116 to remove the alternating component at frequency 2w to leave a detector signal Vd given by: Vd = V 2 1+k 2 sin(-)/2. The detector signal Vd is used as the control voltage for a voltage controlled oscillator (VCO) 118 which controls the up/down direction of the counter 100 by a signal on line 120 and provides count pulses on line 122. The VCO 118 is arranged such that when Vd is greater than zero, then the output value  is increased, and when Vd is less than zero, the output value  is decreased and thus the circuit tends to minimise Vd so that sin(-) tends to zero. The digital output value  is obtained from the counter 100 via a latch 124 and 3-state output buffers 126. Tracking AC analogue to digital conversion devices as illustrated in Fig. 4 are commercially available (for example the model AP 1304 converter or the model 2S56 converter available from Analog Devices, One Technology Way, PO Box 9106, Norwood, Massachusetts 02062, USA), but they suffer from the problem that the gain in the tracking loop varies as a function of the frequency of the modulation component of the gauge signal applied to terminals 70, 72. Fig. 5 illustrates in curve A a typical transfer function H()=/ of such a circuit. To meet the requirements of a metrological instrument capable of measuring both surface texture and form, it is highly desirable that the gain of the analogue to digital conversion circuit should remain within plus or minus 1% of a constant value for a range of frequencies of the modulation component from zero to about 300 Hz. As can be seen in Fig. 5, the curve A fails to meet this requirement and, accordingly, Fig. 5 shows that the known tracking loop type of AC analogue to digital converter is, per se , quite unsuited for use in the surface sensor arrangement of the type of metrological instrument with which the invention is particularly concerned. Fig. 5 also illustrates at curve B, a transfer function F() = ψ/ which would be, in effect, ideal. The preferred embodiment of the invention includes compensating means, indicated at 300 in Fig. 3 for compensating for the transfer function of curve A. The compensating means 300 comprises a microprocessor 302 which receives the digital output from the tracking converter 40 and a memory 304 coupled to the micro-computer 302. In operation, the microcomputer 302 reads, as inputs, a series of digital outputs from the tracking converter 40, stores a predetermined number of these input values in the memory 304, calculates output values ψ and stores a predetermined number of the output values in the memory 304, the calculation of the current output value being dependent on the current and stored input values and the stored output values. More specifically, the calculation is based on the following. The transfer function H() = / of the tracking AC analogue to digital converter is represented in figure 6 from which it will be appreciated that: H() =   = 1+sT1 1+sT1+s 2 /Ka+s 3 T2/Ka where s is the laplacian function jw, T1 and T2 are time constants, typically 2.412ms and 0.345ms, respectively, for the above-mentioned converter, and Ka is a constant, typically 454821s -2 for the above mentioned converter. The desired transfer function F() for the converter 40 and compensating means 300 in combination, as shown by curve B in figure 5, can be represented by: F() = ψ  = wo 2 s 2 +2ρ.wo.s+wo 2 where wo is the desired new cut-off angular frequency, such as 2π.300 radians/s, and ρ is a damping factor. Therefore, the transfer function G() required of the compensating means 300 is given by: G() = ψ/ = F()/H() as illustrated schematically in figure 7. The implementation of the compensation transfer function G() by the microcomputer 302 and memory 304 requires the transformation of the continuous Laplace representation into a discrete representation. This is carried out, for example, by applying a bilinear transformation technique to the G() function to obtain a discrete model of the form: ψ  = a 0 .Z 0 +a 1 .Z -1 +a 2 .Z -2 +....+a n .Z -n b 0 .Z 0 +b 1 .Z -1 +b 2 .Z -2 +....+b n .Z -n where ψ represents the output from the compensation means, a 0 , a 1 ... a n , b 0 , b 1 ...b n are constants, and Z 0 , Z -1 , Z -2 ....Z -n represent the current, last but one, last but two, ... last but n, sampling instants in the operation of the compensation means, respectively, so that Z 0  represents the current input to the compensation means, Z -1  represents the last but one input to the compensation means, Z 0 ψ represents the current output from the compensation means, and so on. For further explanation of the above and of the generation of the constants for a desired transfer function, reference is directed to Forsythe W, "A new method for the computation of digital filter coefficients", Simulation 44:1 23-31 January 1985. It should be noted that it is permitted to treat Z 0 , Z -1 etc as separate mathematical entities, and their superscripts 0, -1, etc can be treated as powers. This has the consequence that Z 0 equals 1, as any value to the power of zero equals one. The above equation can be expressed alternatively in the following way: Z 0 ψ = (1/b 0 ) {[a 0 .Z 0  + a 1 .Z -1  +....+ a n .Z -n ] -[b 1 .Z -1 ψ + b 2 .Z -2 ψ +....+ b n .Z -n ψ]} Thus it can be seen that the current output value Z 0 ψ from the compensating means 300 is an arithmetic function of the current input value Z 0 , the last n input values Z -1  to Z -n  , and the last n output values Z -1 ψ to Z -n ψ. The compensating means 300 may be arranged to perform this calculation as a recursive filtering operation using as few as, for examples, the last five input and output values, that is to say n=5. However, alternatively, non-recursive or finite impulse response techniques may be used, and in this case, for example, the last seventy-nine input and output values may be used. In view of the amount of processing which is then involved, the compensating means is preferably implemented using a finite impulse response (FIR) digital filter device, such as device DSP56200 available from Motorola Inc, USA. The FIR filter is shown schematically in figure 8. The current input value Z 0  is input from the microcomputer 302 to a shift register 400, which stores the current and last seventy-nine input values. A further shift register 402 stores the last seventy-nine output values. As shown, each of the stored input values and output values is multiplied by the respective coefficient a 0 to a 79 and -b 1 to - b 79 , and the products are summed. The resulting sum is divided by the coefficient b 0 to produce the current output value Z 0 ψ which is then returned to the microcomputer 302 and is also input to the shift register 402. The coeffients a 0 to a 79 and b 0 to b 79 are downloaded from the microcomputer 302 during initialisation of the apparatus. In Fig. 5, curve C illustrates the resulting gain of the converter after compensation obtained in an experimental circuit. As can be seen, curve C remains within the plus or minus 1% limit until a point very close to 300 Hz. As noted above, analogue to digital conversion devices as shown in Fig. 4 are commercially available and provide highly stable operation. In view of the 15 bit output, a range to resolution ratio of approximately 32,000 is available from such a circuit. However, the range of displacements of the transducer over which it provides an output with good linearity is typically too small to match this range to resolution ratio. The linearity of the output will vary depending on the precise nature of the transducer used. However, a small size transducer may perhaps have an output error of 0.1% at displacements of the core of 0.1mm in either direction from a centre position, which would be its normal operating range, whereas the range-to-resolution ratio given above may require it to move by up to 0.5mm from the centre position, at which displacement the error may have risen to 5%. Such a transducer cannot provide the high range-to-resolution ratio with an acceptable level of accuracy over the full range. The non-linearity with large core displacements can arise through a variety of factors. Typically, movement of the stylus 19 causes the stylus arm 18 to pivot about a pivot point between the stylus 19 and the core 52. The consequent arcuate movement of the core 52 causes it to change its angle with respect to the coils 56, 58 as it moves. Additionally, large displacements will move the core 52 laterally so as to vary the spacing botween it and the coils 56, 58. Thus, large displacements result in non-linear changes in the coupling between the core 52 and the coils 56, 58, and these changes will not have matching effects on the two coils 56, 58. The arrangement of coils 56, 58 shown in Figure 3 enables it to act as a differential transducer, in which movement of the core 52 results in inductance being lost from one coil and gained by the other. However, with large movements the core 52 can become only very weakly coupled to one of the coils, so that the changes in inductance of the two coils no longer match. Further inaccuracies may arise from the finite size of the tip of the stylus 19. A preferred aspect of the present invention compensates for this non-linearity, so as to extend the range of displacements over which the transducer 16 can be used, in order to enable the high range to resolution ratio available from the digital-to-analogue converter to be used accurately. In order to carry out this compensation, the computer 46 is programmed to obtain a signal Z representing the true displacement of the core 52 (and therefore the true displacement of the stylus tip in response to features of the workpiece surface) by performing the following computation on the signals now represented as z obtained by microprocessor 42 from the output of device 40 and stored in memory 44: Z = Az + Bz 2 + Cz 3 In this equation, A, B and C are calibration constants. These are calculated, before operation of the machine, in a calibration operation in which, as shown in Fig. 9, the stylus 19 is caused to traverse the surface of a calibration ball 500 from a position 502 through a centre position 504 to a final position 506 which are such that, during this movement, the stylus moves throughout its entire range of operation. It will be noted from Fig. 9 that the stylus 19 is mounted for pivotal movement about a pivot point 508 and as a result movement in the Z direction (vertical as seen in Fig. 9) is accompanied by a small amount of movement in the X direction (horizontal as seen in Fig. 9), as will be appreciated from a comparison of the full and broken line positions of the stylus 19 in Fig. 5. The microprocessor 42, in addition to causing the z values of the transducer output to be stored in memory 44, is effective to store therein the x position of the stylus as indicated by counting the pulses which occur on line 39. However, due to the movement of the stylus tip in the X direction which accompanies pivoting of the stylus about 508, these x values as initially stored in memory 44 are incorrect. In the past, where such x direction errors have been compensated for, this has been done by adjusting the speed of traverse of the stylus or the speed of movement of a recording instrument (see e.g. US-A-4 736 208). This is not entirely satisfactory, and does nothing to provide a signal having corrected x values. In the illustrated embodiment, the x values as stored in the memory 44 are corrected. Computer 46 achieves this by carrying out the following computation: X = x + Dz + Ez 2 X is the corrected x value In the above equation, D and E are also calibration constants. The constants A, B, C, D and E in the above equations are calculated during the calibration operation illustrated in Fig. 9 in the following iterative manner. Step 1. Firstly, the stylus 19 is caused to traverse the calibration ball 500 of known radius R and N samples of stylus signal zi (i = 1, 2, ..., N) and of the corresponding x position xi are taken and stored. Step 2. From the values z1, x1 at the initial position 502, z(N/2), x(N/2) at the centre position 504, and zN, xN at the final position 506, the approximate centre coordinates (a, b) of the calibration ball 500 are determined. Step 3. Initial values of the calibration constants are then taken to be A=1, B=0, C=0, D=0, E=0. Step 4. With these calibration constants, corrected values Zi, Xi of the samples are determined, and a series of evaluation values Ti is calculated as follows: Zi = Azi + Bzi 2 + Czi 3 i = 1 to N Xi = xi + Dzi + Ezi 2 i = 1 to N Ti = {R 2 - (Xi-a) 2 - (Zi-b) 2 }/2 i = 1 to N Step 5. Increments dA, dB, dC, dD, dE, da and db of the calibration constants A to E and of a and b, respectively are chosen so as to minimise a sum of squares difference measure F, where F =ΣFi 2 i = 1 to N, and Fi = Ti - (Xi-a)da - (Zi-b)db - (Zi-b)zidA - (Zi-b) zi 2 dB - (Zi-b)zi 3 dC -(Xi-a)ZidD - (Xi-a)Zi 2 dE Step 6 . The calibration constants and a and b are then incremented as follows: new (A) = old (A) + dA new (B) = old (B) + dB new (C) = old (C) + dC new (D) = old (D) + dD new (E) = old (E) + dE new (a) = old (a) + da new (b) = old (b) + db Step 7. It is then determined whether the total of the absolute values of the corrections made is less than a predetermined tolerance value tol, that is abs(dA)+abs(dB)+abs(dC)+abs(dD)+abs(dE) < tol? If not, then steps "4" to "7" are repeated using the revised calibration constants new (A) to new (E), and new (a) and new (b), but if so the calibration constants are stored in the memory 44 (preferably a non-volatile part thereof) for use in subsequent measuring processes, and the calibration operation is terminated. It will be apparent that various modifications can be made to the calibration process described above. For example, steps "4" to "6" may be carried out, say, six times in succession before step "7" is performed for the first time, and also if, say, twenty iterations do not provide a correction within tolerance, the calibration process may be terminated and a warning message may be given regarding non-convergence of the process. The calibration technique described above employs a solution to a non-linear least squares problem using the Gauss-Newton algorithm and a QDR decomposition algorithm employing Givens transformations. In this connection, references is directed to Kennedy, William J., Jr.; Gentle, James E. "Statistical Computing", (1980), Marcel Dekker Inc. It will be apparent, however, that other standard non-linear least squares problem solving techniques may be applicable. Utilising this technique in an experimental apparatus, a range of 1.18 mm with a resolution of 37 nm, giving a range to resolution ratio of about 32000, has been achieved with an accuracy of 200 nm. Since the experimental apparatus involves an inductive transducer as described with reference to the drawings, this represents a substantial improvement in the state of the art. Various other modifications are possible within the scope of the invention. For example, range switching may be included and, if desired, the compensation represented by the equation Z = Az + Bz 2 + Cz 3 may be selectively applied according to the selected range. For example, in certain ranges no compensation, or a different form of compensation, such as represented by the Z = Az + Bz 2 may be applied.

Claims: A metrological apparatus for traversing a stylus (19) across a surface (6) to obtain an electrical transducer signal representative of a characteristic of the surface (6), the apparatus comprising a stylus (19) for contacting a surface (6), a drive (20) for traversing the stylus (19) across the surface (6) in a first direction (X), means (26 to 38) for providing a first position signal in accordance with the position in the first direction (X) of the point of contact of the stylus with the surface, and a transducer (16) responsive to movement of the stylus (19) in a second direction (Z) towards and away from the surface (6) for producing the electrical transducer signal representative of a surface characteristic, the position of the point of contact in the first direction (X) varying with its position in the second direction (Z), characterised in that the apparatus further comprises processing means (46) for outputting a compensated signal, the value of which represents the position of the said point of contact in the first direction (X) when compensated for said variations in the position of the point of contact in the first direction (X) arising from variation in its position in the second direction (Z), the processing means calculating the value of the compensated signal from the first position signal and the transducer signal. An apparatus according to claim 1 which comprises memory means (44) for storing a series of values of the said compensated signal and a series of transducer signal values. An apparatus according to claim 1 or claim 2 in which the stylus (19) is mounted by a pivot (508) which is offset in the first direction (X) from the point of contact, and the said variation in the position in the first direction (X) of the point of contact with its position in the second direction (Z) comprises variation in the distance in the first direction (X) between the position of the point of contact and the position of the pivot (508) due to pivoting of the stylus, the first position signal represents the position of the pivot (508) in the first direction, and the processing means (46) calculates the value of the compensated signal so as to be compensated for variations in the said distance. An apparatus as claimed in any one of claims 1 to 3, wherein the processing means (46) is bperable to use the transducer signal and a calibration constant D in producing the compensated signal. An apparatus as claimed in any of claims 1 to 4, wherein the processing means (46) is operable to use the square of the transducer signal and a calibration constant E in producing the compensated signal. A metrological apparatus according to any one of claims 1 to 5 in which the transducer (16) has a normal range over which the electrical transducer signal is substantially linearly related to a position detected by the transducer and an extended range beyond the normal range, over which the electrical transducer signal is non-linearly related to a position detected by the transducer, and the said processing means (46) processes the said electrical transducer signal to compensate for non-linearity over the extended range so that the compensated signal is substantially linearly related to the detected position over the normal and extended ranges of the transducer. An apparatus as claimed in claim 6, wherein the processing means (46) is operable to use the square of the transducer signal and a calibration constant B in producing the compensated signal. An apparatus as claimed in claim 6 or claim 7, wherein the processing means (46) is operable to use the cube of the signal and a calibration constant C in producing the compensated signal. An apparatus as claimed in any preceding claim, wherein the transducer (16) comprises a pair of symmetrical coils (56, 58) and a core (52) movable relative to the coils, the coils being connected as two arms of a bridge circuit. A method of calibration of an apparatus as claimed in any of claims 4, 5, 7 and 8, comprising the steps of performing a traverse with respect to a surface (500) of known form, storing a series of the electrical signals obtained during the traverse, setting an initial value for the calibration constant or each of the calibration constants, and determining a correction to the calibration constant or each of the calibration constants so that the stored signals as compensation processed using the corrected calibration constant(s) tend to equal the signals expected from the surface of known form by an apparatus providing fully compensated signals. A method as claimed in claim 10, wherein the correction determining steps includes the step of calculating a difference measure for each stored signal involving a variable correction to the calibration constant or each of the calibration constants, and using a least squares solution to determine values of the corrections which minimise the sum of the squares of the difference measure. A method of measuring a surface characteristic of a workpiece (4), comprising: traversing a stylus (19) across a surface (6) of the workpiece (4) in a first direction (X); reading an electrical signal representative of the surface characteristic from a transducer responsive to movement of the stylus (19) in a second direction (Z) towards and away from the surface (6); and reading a first position signal provided in accordance with the position in the first direction of the point of contact of the stylus (19) with the surface (6), the position of the point of contact in the first direction (X) varying with its position in the second direction (Z), characterised in that the method comprises generating a compensated signal, the value of which represents the position of the said point of contact in the first direction (X) when compensated for said variations in the position of the point of contact in the first direction (X) arising from variation in its position in the second direction (Z), the value of the compensated signal being calculated from the first position signal and the transducer signal.

Inventors: BUEHRING IAN KARL, GARRATT JOHN DAVID, SCOTT PAUL JAMES, BUEHRING, IAN KARL, GARRATT, JOHN DAVID, SCOTT, PAUL JAMES
Applicants: RANK TAYLOR HOBSON LTD, RANK TAYLOR HOBSON LIMITED
Application year: 1990
Main classification: G01R15
IPC codes: G01B7