Reset pulse circuits

Abstract: A power-on reset circuit provides an automatic reset pulse immediately after power-up. The supply voltage is provided to the reset circuit. An RC filter (90) with a variable time constant provides a "hump" waveform which is coupled to a waveform shaper (91). The waveform shaper converts the hump voltage output from the RC filter into a reset signal. Latching mechanisms (92) are included to prevent refiring of the reset signal.

Description: This invention relates to reset pulse circuits. In complex electrical circuit designs, it is often necessary on power-up to set certain circuit nodes or circuit elements to a known state. For example, when the ON switch is enabled in computer circuits, the computer clocks and other counters need to be initiated, started or reset. Power-on reset circuits are also often required for dynamic random access memory (DRAM) operation. The reset function is most easily accomplished by an automatic pulse which executes immediately after power is turned on. The automatic reset circuit should only act immediately after power-up; accidental firing unnecessarily interrupts the use of the circuit. An effective reset circuit should not inadvertently misfire. US 4365174 discloses a circuit which may be used for providing a reset pulse signal having an input terminal for receiving a signal from a pulse generator. Means for providing a time delay between the applied power and the termination of the reset pulse comprises a charge sharing mechanism including two capacitors, two switches and a threshold detector invertor. In EP 0206084, a charging current is generated in a capacitor (CO in Fig 1) when the supply potential is ramped up. The generated current is then mirrored to an MOS device (Q4 in Fig 1) through two cascaded current mirror stages. The time at which the generated reset pulse terminates is controlled by the magnitude of the charging current and the multiplication factor of the current mirror stages. In US 3895239 the pulse generated by the reset circuit is primarily controlled by the threshold voltages of P-MOS transistor 24 and the forward-biased turn-on voltage of diode 20, both shown in Fig. 1, in conjunction with the ratioed-inverter consisting of devices 28 and 30 (see lines 64-68 column 3 and lines 9-20 column 4). This circuit relies on the time period during which VCC travels from one Vt to two VTs to provide the time delay between the application of power and the termination of the reset pulse. EP 0323367 discloses a circuit for providing a reset pulse signal upon power-up, the circuit comprising: receiving means in the form of FETs D1, D2 and T8; pulse shaping means in the form of FETs T4 and T5; and feedback means in the form of a circuit C. Prior art preset circuits are electronically cumbersome because they use discrete components to achieve the characteristics. The present invention is a device to perform the automatic reset function needed in electrical circuits, which it does with greater efficiency. It is not dependent on discrete resistive and capacitive components that reside off-chip. The present invention eliminates external power-on circuits, prevents accidental resetting, and is more simple than prior art designs. According to the present invention, there is provided a circuit coupled between a power supply terminal receiving a supply voltage and a ground terminal receiving a reference potential for providing a reset pulse signal upon power-up, the circuit comprising: receiving means having an input terminal for receiving a feedback signal and an output terminal, wherein the receiving means provides on said output terminal a signal having a voltage level intermediate the supply voltage and the reference potential; charging and discharging means comprising a coupling capacitor and a controllable and variable resistance, the charging and discharging means having a first input terminal coupled to the output terminal of the receiving means, the first input terminal receiving the signal on the output terminal of the receiving means for controlling the controllable and variable resistance, a second input terminal, and an output terminal for providing an output signal; pulse shaping means having an input terminal coupled to the output terminal of the charging and discharging means, and an output terminal for providing the reset pulse signal, wherein upon the supply voltage increasing from the reference potential, the circuit generates the reset pulse signal which terminates at a time substantially controlled by the charging and discharging means; and feedback means having an input terminal for receiving the reset pulse signal, and having an output terminal for providing the feedback signal, the output terminal being coupled to the input terminal of the receiving means and the second input terminal of the charging and discharging means, wherein the feedback signal disables the receiving means and the charging and discharging means when the feedback signal assumes any one of a selected range of values upon termination of the reset pulse signal. The present invention provides a single pulse reset circuit to set critical circuits to a known state during power-on. This permits proper circuits operation. A supply voltage Vcc is not an "instant on" signal, but has a rising edge which ramps up to full amplitude. The reset circuit accepts the supply voltage Vcc and provides a single pulse output "PONRST" at the rising edge of Vcc. The reset circuit of the present invention is implemented in an integrated fashion with no discrete devices required. The reset circuit is less susceptible to unintentional firing due to power source transients or other interferences. The reset circuit consists of a number of subsystems. An intermediate level reference generator outputs a voltage signal whose level is approximately centred between the supply voltage waveform and 0V. This intermediate level reference voltage is used to control a transistor that functions as a variable resistor in a subsequent circuit block. With this variable resistor, an RC element within a charging and discharging block has a variable time constant. The time constant, along with the resistance, starts at infinity and decreases to measurable values during operation of the circuit. The output of this charging and discharging block is a "hump" waveform, which feeds into the next sequential circuit block, a pulse shaper. The pulse shaper converts the hump voltage output from the charging and discharging block into a signal which mirrors the waveform of the supply voltage for a short interval of time. The pulse shaper is buffered so that it may be connected to other circuits, where the power-on reset pulse is needed. The power-on reset pulse is fed back through feedback lines and feedback block to disable the intermediate level reference generator and the charging and discharging block upon termination of the reset pulse. The circuit of the present invention is thus able to perform the automatic reset function required of single pulse-type power-on reset signals, that is to set critical circuit components to known states during power-on to guarantee the proper operation of a circuit design. The circuit of the present invention moreover self-protects from power source transients and glitches, and does not need to use discrete electrical components, that is, it is an integrated circuit implementable design. The invention is further described below, by way of example, with reference to the accompanying drawings, in which: Figure 1 is a schematic representation of a reset pulse circuit in accordance with the present invention, Figure 2 is a block diagram of the circuit of Figure 1, Figure 3 is a timing diagram for various nodes of the circuit, Figure 4 is a detailed view of the timing diagram of Figure 3, and Figure 5 is a flow diagram of the operation of the circuit of the present invention. In the following description of a CMOS power-on reset circuit embodying the invention, numerous specific details are set forth such as voltage polarity, semi-conductor type, etc., to provide a more thorough understanding of the present invention. It will be obvious however, to one skilled in the art, that the present invention may be practiced without these specific details. In other instances, well-known circuits have not been described so as not to obscure the present invention. The present invention is a circuit which provides an automatic reset pulse immediately after power-up. The only input to the circuit is the power voltage, V cc . The output of the circuit is a signal called PONRST. The reset circuit is comprised of an RC filter with dynamic time constant, a waveform shaper, an enable and latch block, and intermediate-level reference generator. These circuit blocks work sequentially to provide a single reset pulse at start-up. Start-up is characterized by the supply voltage, V cc , rising from 0V in a ramp waveform up to full level, V cc . The present invention provides a signal, PONRST, which closely follows the V cc signal while it is in the ramp stage and continues following V cc for a short interval of time after it has reached full voltage. After this interval, PONRST drops to 0V and does not fire again. Figure 2 is a block diagram of the present invention. V cc is input to all four functional blocks. RC filter 90 receives V cc at input 10, V A at input 14, and V B at input 15. RC filter 90 outputs signal V C at its output 19. Pulse shaper 91 receives as input V cc at input 11, and the signal V C at input 16. Pulse shaper 91 outputs signal V D at its output 20. V D feeds through inverters I1 and I2 to generate the signal PONRST. The effect of inverters II and I2 is to buffer the signal power level between the present invention and circuits to which it is attached. V D feeds into feedback block 92 at input 17. Feedback block 92 also receives V cc at input 12. Feedback block 92 outputs signal V B on output 21. Intermediate level reference generator 93 receives V cc at input 13, and V B at input 18. Intermediate level reference generator 93 has output signal V A on its sole output 22. Charging and discharging block 90 is composed of transistors and capacitors. The transistors operate as switches or variable resistors. Charging and discharging block 90 receives as input V cc , V A ,and V B . Charging and discharging block 90 outputs a signal labeled V C . At a functional level, V C , the output of charging and discharging block 90 is dependent upon inputs V cc , V A , and V B . At power-up, V C closely follows the waveform of V cc . Subsequently, when V A or V B reaches a certain voltage (the turn-on voltage, V T of charging and discharging block 90 internal transistors), V C begins to drop to 0V. Going up, V C is a ramp function because the input signal V cc , which V C follows, is a ramp function. Also, when V A or V B goes high, V C , the output of charging and discharging block 90, ramps down. At power-up, V C , the output of the charging and discharging block 90, closely follows V cc input. The accuracy of signal V C 's approximation of V cc depends upon capacitor internal to RC filter 90. The internal capacitor, along with internal transistors which act as variable resistors, provide an RC element with a variable time constant. During initial power-on, the time constant is almost infinite because internal transistors are set at infinite resistance. During the operation of the circuit, that resistance, and hence the time constant associated with it, drops from infinity toward measurable values. As this happens, the time constant also drops. Pulse shaper 91 executes its function upon a signal from charging and discharging block 90, signal V C . V C , as mentioned earlier, closely follows V cc out to a certain value. Pulse shaper 91, which receives V C as an input, turns on its output V D when V C reaches a certain level (the turn-on voltage, V T , of transistors internal to the pulse shaper 91). When this turn-on effect occurs, V D follows V cc even more closely than V C . In fact, V D , after being turned on, is exactly the same value and shape as V cc . The output of pulse shaper 91, V D , is dependent on the value of the input V C . When V C rises beyond a certain voltage, V D closely follows the other input to pulse shaper 91, V cc . Conversely, when V C drops below that certain value, then signal V D , the output of the pulse shaper 91, is set to 0V. The output V D of pulse shaper 91 is an input to feedback block 92. The feedback block 92 is essentially an inverter and clamp. When V D , the input to this function block is high, the output, V B , goes low and is held low. When V D , the input to block 92 goes low, V B goes high. V B , the output of the feedback block 92, is coupled to the intermediate level reference generator 93 and charging and discharging block 90. Intermediate level reference generator (ILRG) 93 receives V cc and V B as input, and has one output, V A , which is one of the inputs to charging and discharging block 90 discussed earlier. ILRG 93 sets its output, V A , dependent on its input, V B . When the input V B is clamped to 0V, V A begins ramping up in voltage. The slope of the ramp of V A is less than the slope of V C , V D , and V cc , which are ramp functions as well. When input V B to ILRG 93 reaches a certain voltage (the turn-on voltage, V T , of transistors in this circuit block) then the output, V A , drops off to 0V very quickly. V A is unlike signal V C which ramps up in voltage and then ramps down. V A , although it ramps up, drops off abruptly to 0V. The behavior of the various signals of the reset circuit is illustrated in Figures 3 and 4. Figure 3 is a timing diagram of voltage versus time. Figure 4 is a timing diagram of voltage versus time for the circuit signals at the early stages of operation of the reset circuit. Referring first to Figure 4, at time T 0 , V cc is turned on and begins to ramp up to full voltage. V C follows V cc very closely and instantaneously at time T 0 . V A and V B each remain low, charging capacitors Q3 and Q17, respectively. V D is also low immediately after V cc is turned on. At time T1, V C has reached a voltage sufficient to turn on transistor Q7 which brings V D up to approximately the level of V cc . At this point, V D turns on Q15, clamping V B to 0. With V B clamped to 0, V A begins ramping up in voltage, charging capacitor Q3 if V cc is greater than |VTP(Q1)| + |VTP(Q2)|. Referring now to Figure 3, signals V A , V C , V D and V cc all continue ramping upward until time T 2 . At time T 2 , V cc and V D reach their peak values and level off. At time T 3 , the voltage at V A has reached the threshold voltage of transistor Q5. This causes V C to begin to discharge slowly through transistor Q5. V A continues increasing until time T 4 when it reaches a peak value, approximately one-half that of V cc . At time T 5 , V C falls below the threshold voltage of transistor Q7. This causes V D to discharge. Accordingly, PONRST falls as well. At time T 6 , V D falls below the threshold voltage of transistor Q14, allowing V B to begin to rise again at time T 6 . At time T 7 , voltage V B has reached a level sufficiently high to latch V A and V C to 0 volt through transistors Q19 and Q16, respectively. At this time, transistor Q16 is in the triode region. At time T 8 , V D has reached 0 volt. At time T 9 , V A has reached its low voltage point. At time T 10 , V B has risen to approximately V cc . As a result, node C is damped at 0 volt through transistor Q16 regardless of power supply surges which might couple V C high or low after the initial reset pulse. When V B is high, the intermediate level reference generator is disabled such that no standby current is generated after the reset function expires. Referring to Figure 3, one can see that V D provides a unitary reset pulse at power-up. This reset pulse closely follows V cc , and is very stable after its initial firing. Stability is important because subsequent inadvertent firing of the reset is very disruptive to microprocessor and microprocessor-dependent machines. Figure 5 is a decisional flow chart of the operation of the present invention. Various internal turn-on voltages are able to execute logic-like functions and enable the circuit to decide when to execute a function or to bypass that function. In Figure 5, begin block 1 corresponds to T = 0. At execution block 2, the power switch has been turned on and V cc begins its ramp function up to full voltage. In decision block 3, there is a check to see if V A is greater than the turn-on voltage of transistor Q5. In the early stages of this circuit, when V cc has just been turned on, there will be no appreciable voltage at V A , and the function of the circuit flows into decision block 4. Decision block 4 likewise does a voltage level check by comparing V B to the turn-on voltage of transistor Q16. Again, because voltage is insufficient, this decision will be negative, i.e., V B will be less than the turn-on voltage of transistor Q16. Next, execution block 5 turns on the ramp function of V C . The chart flows into decision node 6 where it checks to see if V C has increased above a certain level, a turn-on voltage of Q7. If not, the logic cydes back into execution block 5 where V C continues ramping. The flow chart cydes between execution block 5 and decision block 6 until the condition in decision block 6 becomes affirmative and the flow continues into execution block 7. At that point, V D jumps up to the V cc level. Following execution block 7, decision block 8 checks to see if node V D voltage is greater than the turn-on voltage of transistor Q15. If not, the flow cycles back to execution block 7 where V D again is forced toward V cc . Finally, when the condition of decision block 8 is satisfied, the circuit continues into execution block 9. The voltage at node V B is driven toward 0V. In decision block 10, if V cc is not greater than the turn-on voltages of transistors Q1 and Q2, then the circuit returns to execution block 9 and V cc continues ramping up toward its final value. When the condition at decision node 10 is satisfied, then the circuit proceeds to execution block 11 and V A begins a ramp function. Because as soon as V cc reaches voltage level greater than |VTP(Q1)| + |VTP(Q2)|, the ILRG will be activated provided that V B is lower than V cc - |VTP(Q18)|. From execution block 11, the circuit passes through merge node 21 and returns to decision block 3. Supposing that V A is still below the turn-on voltage of transistor Q5 (the condition of decision block 3), then the circuit proceeds again through decision block 4. V B has already been forced to 0V so the decision at block 4 is negative. At execution block 5, V C continues ramping. At node 6, nothing has changed so that the condition in decision block 6 is still satisfied, i.e., V C is greater than turn-on of transistor Q7. At execution block 7, V D is at V cc so the circuit flows through execution block 7 and decision block 8 without pausing. The execution of block 9 and the decision of block 10 are already achieved so that the circuit flows through the states represented by these blocks without hesitating. At execution block 11, V A continues ramping and the circuit cycles back up to decision block 3. Now, supposing that V A is greater than the turn-on voltage of transistor Q5, the circuit proceeds to execution block 14. V C begins to deramp. In decision block 15, if node V C is greater than the turn-on voltage of transistor Q7, then the circuit returns to execution block 14 and V C continues deramping. Once the condition in decision block 15 is satisfied, the circuit proceeds to execution block 16. In that block, V D is forced to 0V. In decision block 17, there exists a condition that if V D - V cc is not less than the turn-on voltage of transistor Q14, then the circuit returns to execution block 16 and V D is again forced to 0V. Once the condition of decision block 17 is satisfied, the circuit continues to execution block 18. In execution block 18, V B is forced to V cc . In decision block 19, if V B has not yet passed the turn-on voltage of transistor Q19, then the circuit returns to execution block 18. Once the condition in decision block 19 is satisfied, the circuit continues to execution block 20 and V A is forced to 0V. From execution block 20, the circuit follows through merge node 21 and back to decision block 3. V A , which has just been forced toward 0V, may still satisfy the condition of decision block 3. Therefore, the circuit will again cyde to execution block 14, and V C will continue deramping. Circuit function flows into decision block 15, whose condition was satisfied in the previous pass. The circuit advances to execution block 16 which has already been executed, and node 17 whose condition also has been satisfied. The circuit continues to execution block 18 which has already been executed, and whose condition, contained in decision block 19, has also been satisfied. The circuit continues again to execution block 20 and V A is continued to be forced to 0V. The circuit flows again through merge node 21 and back to decision block 3. Supposing now that V A , forced to 0V in execution block 20, is below the turn-on voltage of transistor Q5, the circuit continues downward into decision block 4. Unlike the first pass through this decision block, V B is now at V cc level (look back to execution block 18). Therefore, the circuit state flows into execution block 12 and V C is forced to 0V. The circuit state then flows to decision block 13 and if V C is not below the turn-on voltage of transistor Q7, then V C is forced to 0V again. When the condition at decision block 13 is satisfied, i.e., V C is less than the turn-on voltage of transistor Q7, then the circuit flows back into execution block 16. V D is again clamped to 0V. By the action of execution block 16, the condition in decision block 17 is satisfied and the circuit flows again to decision block 18, where V B is clamped to V cc level. By the action of execution block 18, the condition of decision block 19 is satisfied and the circuit flows into execution block 20. In execution block 20, V A is again clamped to 0V. The final state of the circuit can be traced through a path of the logic flow diagram, through which the circuit in its final state cycles. The circuit begins at decision node 3, because V A is now clamped at 0V, and the circuit flows to decision node 4. Because V B is now at V cc level, the circuit flows through execution block 12 and decision block 13. Because V C is damped at 0V, the circuit Rows through these blocks without hesitation. Similarly, because V D is already at 0V, the circuit flows through execution block 16 and decision block 17 without pausing. V B is already at V cc level. Therefore, execution block 18 and decision block 19 are sped through. Finally, because V A is at 0V, there is no need to execute the function of execution block 20. The circuit finds its final state by continually cycling through these blocks: 3, 4, 12, 13, 16, 17, 18, 19, 20 and 21, and then back to 3. This effect a stable state because none of the execution blocks or decision blocks will now effect any change in the operation. Figure 1 is an electrical schematic of the present invention. It illustrates N-channel and P-channel field-effect transistors, integrated circuit capacitors, resistors, and inverters. Figure 1 divides the circuit into the component subcircuits. Charging and discharging block 90 comprises P-channel capacitor Q4, N-channel capacitor Q3, and N-channel transistors Q5 and Q16. Capacitor Q4 is connected between V cc input 10 and output node 19 of charging and discharging block 90. Capacitor Q3 is connected between input node 14 of charging and discharging block 90 and ground. Transistors Q5 and Q16 are parallel connected, drain-drain and source-source, between output node 19 of charging and discharging block 90 and ground. Drain 50 of transistor Q5 is connected to drain 51 of transistor Q16 which is connected to output node 19. Gate 52 of transistor Q5 is connected to input node 14. Gate 53 of transistor Q16 functions as input 15 to charging and discharging block 90. Source 54 of transistor Q5 and source 55 of transistor Q16 are common-connected to ground. Output node 19 is connected to interconnect line 40, which transmits the output signal of charging and discharging block 90 to the pulse shaper 91. Input node 14 is connected to line 44, from which it receives an input signal, V A , to charging and discharging block 90. Gate 53 of transistor Q16, which performs the function of input 15 for RC filter 90, is connected to line 42, on which it receives input signal V B from feedback block 92. Line 40, the carrier of V C , connects output 19 of charging and discharging block 90 to input 16 of pulse shaper 91. Pulse shaper 91 consists of complementary transistor pairs Q6-Q7, Q8-Q9, Q10-Q11 and Q12-Q13. Q6, Q8, Q10 and Q12 are P-channel transistors and Q7,Q9,Q11 and Q13 are N-channel transistors. The complementary pairs have particular width-to-length (W/L) ratios. The W/L ratios for these transistors are as follows: Q6/Q7 is approximately 1:5 Q8/Q9 is approximately 15:1 Q10/Q11 is approximately 1:5 Q12/Q13 is approximately 2.5:1 Sources 61, 67, 73 and 79 of transistors Q7, Q9, Q11 and Q13, respectively, are common-connected to ground. Sources 56 and 62 of transistors Q6 and Q8, respectively, plus a terminal of resistor R1, function as input 11 of pulse shaper 91 which receives the waveform of V cc . The other end of resistor R1 is connected to sources 68 and 74 of transistors Q10 and Q12, respectively. Input node 16 of pulse shaper 91, which receives signal V C on line 40, is connected to gates 57 and 60 of transistors Q6 and Q7, respectively. Drains 58 and 59 of transistors Q6 and Q7, are connected through line 80 to gates 63 and 66 of transistors Q8 and Q9, respectively. Drains 64 and 65 of transistors Q8 and Q9, respectively, are connected through line 81 to gates 69 and 72 of transistors Q10 and Q11, respectively. Drains 70 and 71 of transistors Q10 and Q11, respectively, are connected through line 82 to gates 75 and 78 of transistors Q12 and Q13, respectively. Drains 76 and 77 of transistors Q12 and Q13, respectively, are connected through line 83 to output node 20. V D , the signal on output line 83 and output node 20, feeds into inverters I1 and I2 connected in series. Output line 84 transmits the power-on reset signal, PONRST. Output node 20 is connected to line 41 which carries the signal V D to feedback block 92 at its input 17. Feedback block 92 consists of P-channel transistor Q14, N-channel transistor Q15 and N-channel capacitor Q17. Transistors Q14 and Q15 form a complementary pair where the W/L of transistor Q14 is very small compared with the W/L of Q15. Input line 41 is received at input 17 of feedback block 92. Input node 17 is connected to gates 85 and 88 of transistors Q14 and Q15, respectively. Source 84 of transistor Q14 is connected to input 12 of feedback block 92 where is received the waveform of V cc . Source 89 of transistor Q15 is connected to ground. Drains 86 and 87 of transistors Q14 and Q15, respectively, are connected together. They are also attached via line 43 to output 21 of feedback block 92. Capacitor Q17 shunts line 43 to ground. Output node 21 is connected to ground through capacitor Q17. The signal on output 21, V B , is connected to charging and discharging block 90 at input 15 via line 42. V B also connects to input 18 of intermediate level reference generator 93 via line 43. Intermediate level reference generator (ILRG) 93 receives the signal on line 43 from feedback block 92 at its input 18. ILRG 93 also receives V cc waveform on input 13. ILRG 93 is composed of three P-channel transistors Q18, Q1 and Q2, and a single N-channel transistor, Q19. Input 18 is connected to gate 110 of transistor Q19 and to gate 101 of transistor Q18. Source 100 of transistor Q18 is connected to input 13. Drain 102 of transistor Q18 connects to source 103 of transistor Q1. Gate 104 of transistor Q1 connects to node 112. Drain 105 of transistor Q1 also connects to node 112. Source 106 of transistor Q2 also connects to node 112. Node 112 connects to output 22 via line 44. N-well of transistor Q2 connects to node 112. Gate 107 of transistor Q2 connects to ground. Drain 108 of transistor Q2 also connects to ground. Drain 109 of transistor Q19 connects to output 22. Gate 110 of transistor Q19 connects to input 18. Source 111 of transistor Q19 connects to ground. Output signal V A is to be found at output 22 and transmitted via line 44 to charging and discharging block 90. At power-on, capacitor Q4 and transistor Q5 function an RC element (charging and discharging block 90). The time constant is almost infinite at power-up because at node 14, the gate of transistor Q5, there is very little voltage and so Q5 is an almost-infinite resistance. For this reason, V C on line 40 dosely follows via capacitor Q4 the waveform of V cc . Pulse shaper 91 is composed of transistors Q6-Q13, resistor R1, and inverters 11 and 12. Inverters I1 and I2 are buffers for the circuits that are to be reset. The P-channel and N-channel ratios of pulse shaper 91 (Q6-Q13) and feedback block 92, (Q14-Q17), are designed such that when V C reaches V T of Q7 (remember, V C is closely following V cc , a ramp function), V D at node 20 and line 41 will quickly jump up and follow V cc . This will, in turn, set V B at node 21 and lines 42 and 43 to 0V, and disable Q16 and Q19. 0V on line 43 turns on Q18 to enable intermediate level reference generator 91 (Q1 and Q2) during the very beginning of power-on. When Q18 is turned on, Q1 and Q2 will form a voltage divider to set V A on line 44 at about 60% of V cc to guarantee low V cc functionality at high V T process corner. Q3 is a capacitor between line 44 and ground which controls the ramping rate of V A . When V B is set to 0V, V A begins ramping up in voltage. As soon as V A on line 44 reaches V T of Q5, V C starts to discharge slowly through Q5. Now charging and discharging block 90, composed of Q4 and Q5, has a time constant which is decreasing from infinity. V A on line 44 continues increasing in voltage until Q5 is forced into a triode mode and V C on line 40 is effectively clamped to ground (time constant = 0). The end result is a "hump" waveform V C created on line 40 during power-on. Resistor R 1 in Figure 1 serves as a current limiter to minimize the switching current during power-on. Q17 is used to hold V B at almost 0V at the very beginning of power-on. Before V A reaches V T of Q5, V C essentially follows V cc and the voltage level of V C on line 40 during this period of time is determined by the coupling efficiency of P-channel capacitor Q4. After V A reaches V T of Q5, the voltage V C on line 40 starts to fall slowly. As soon as V C falls below V T of Q7, V D and the output signal PONRST, will go low and force V B to high via Q14. This will also latch V A and V C at 0V through Q19 and Q16, respectively. At this moment, Q16, connected on line 42 to V B , is in a triode region. V C will be clamped at 0V through Q16 regardless of any power supply surges which might couple V C on line 40 high or low. Also, when V B is at V cc , intermediate level reference generator 93 will be disabled such that no standby current is drawn after the reset function expires. This is the implementation of the no-power characteristic of this device. In summary, the pulse shaper 91 converts the hump waveform of V C into a single-pulse signal PONRST with full V cc swing during power-on. V B serves as the feedback signal through the proper control of Q16, Q18 and Q19 to enable the power-on reset generator and to avoid accidental reset due to power surges. All of these operations are self-timed. No external timing control is required. The minimum reset pulse width is achieved by modulating the turn-on resistance of Q5 to obtain the dynamic RC time constant through intermediate level reference generator 93.

Claims: A circuit coupled between a power supply terminal receiving a supply voltage and a ground terminal receiving a reference potential for providing a reset pulse signal upon power-up, the circuit comprising: receiving means (93) having an input terminal for receiving a feedback signal and an output terminal, wherein the receiving means provides on the output terminal a signal having a voltage level intermediate the supply voltage and the reference potential; charging and discharging means (90) comprising a coupling capacitor and a controllable and variable resistance, the charging and discharging means having a first input terminal coupled to the output terminal of the receiving means, the first input terminal receiving the signal on the output terminal of the receiving means for controlling the controllable and variable resistance, a second input terminal, and an output terminal for providing an output signal; pulse shaping means (91) having an input terminal coupled to the output terminal of the charging and discharging means, and an output terminal for providing the reset pulse signal, wherein upon the supply voltage increasing from the reference potential, the circuit generates the reset pulse signal which terminates at a time substantially controlled by the charging and discharging means; and feedback means (92) having an input terminal for receiving the reset pulse signal, and having an output terminal for providing the feedback signal, the output terminal being coupled to the input terminal of the receiving means and the second input terminal of the charging and discharging means, wherein the feedback signal disables the receiving means and the charging and discharging means when the feedback signal assumes any one of a selected range of values upon termination of the reset pulse signal. A circuit as claimed in claim 1 wherein the receiving means comprises an intermediate level reference generator (93). A circuit as claimed in claim 2 wherein the intermediate level reference generator comprises three field-effect transistors (Q18, Q1, Q2) (FETs) of a first type, and one FET (Q19) of a second type, the three FETs of the first type are connected in series, the FET of the second type is connected in parallel with one of the three series-connected FETs, one of the FETs of the first type (Q18) in combination with the FET (Q19) of the second type enable the intermediate level reference generator when the feedback signal falls below a first voltage level and disable the intermediate level reference generator when the feedback signal exceeds a second voltage level. A circuit as claimed in claim 1, 2 or 3 wherein the feedback means (92) prevents unintentional occurrence of the reset pulse. A circuit as claimed in claim 4 wherein the feedback means comprises an FET (Q14) of the first type, an FET (Q15) of the second type, and a capacitor (Q17) of a first type, the FETs being connected in series between the power supply terminal and the ground terminal, the capacitor being connected between the output terminal of the feedback means and the ground terminal, the reset pulse signal being coupled to the gate terminals of the FETs, and the drain terminals of the FETs providing the feedback signal. A circuit as claimed in any preceding claim wherein the charging and discharging means (90) comprises a capacitor (Q3) of the first type, a capacitor (Q4) of a second type, and two FETs (Q5, Q16) of the second type, the FETs being connected in parallel, the capacitor (Q3) of the first type being connected between the output terminal of the receiving means and the ground terminal, the capacitor (Q4) of the second type being connected in series between the power supply terminal and the drain terminals of the FETs (Q5,Q16) connected in parallel, one of the FETs (Q5) receives at a gate terminal the output terminal of the receiving means, and one of the FETs (Q16) receiving at a gate terminal the output terminal of the feedback means. A circuit as claimed in any preceding claim wherein the reset pulse signal is connected to a first and a second inverter (I1 & I2). A circuit as claimed in any preceding claim wherein the pulse shaping means comprises four complementary pairs of FETs, wherein: the complementary pairs each comprise a FET (Q6, Q8, Q10, Q12) of the first type connected in series with a FET (Q7, Q9, Q11, Q13) of the second type; the complementary pairs provide a first complementary pair output terminal, a second complementary pair output terminal, a third complementary pair output terminal and a fourth complementary pair output terminal, respectively; the complementary pairs are connected in parallel between the power supply terminal and the ground terminal; the output terminal of the charging and discharging means is connected to the gate terminals of a first of the complementary pairs, the first complementary pair output terminal connected to the gate terminals of a second of the complementary pairs, the second complementary pair output terminal connected to the gate terminals of a third of the complementary pairs, the third complementary pair output terminal connected to the gate terminals of a fourth of the complementary pairs, the fourth complementary pair output terminal providing the reset pulse signal; the reset pulse signal is approximately equal to the voltage on the ground terminal when the first output signal on the output terminal of the charging and discharging means is below a first voltage level; and the reset pulse signal is approximately equal to the voltage on the power supply terminal when the first output signal on the output terminal of the charging and discharging means is above the first voltage level. A circuit as claimed in claim 8 wherein two of the series complementary pairs (Q10, Q11 & Q12, Q13) are connected through a resistor (R1) to the power supply terminal.

Inventors: CHANG SHUEN-CHIN, CHANG, SHUEN-CHIN
Applicants: SAMSUNG SEMICONDUCTOR INC, SAMSUNG SEMICONDUCTOR, INC.
Application year: 1990
Main classification: H03K17
IPC codes: H03K17