Magnetic resonance systems

Abstract: A magnetic resonance imaging system includes one or more digital transmitters (B), one or more digital receivers (C), and digital data processing circuitry (D) which are all clocked and controlled by a single clock (F). Each digital transmitter includes a numerically controlled modulated oscillator (20) which processes digital phase and frequency signals to produce an output which addresses a wave-form map stored in a PROM (22). Each wave-form output of the PROM is multiplied (24) by a digital amplitude profile signal to generate phase, frequency, and amplitude modulated digital RF signals. A clock gate (30) controls clocking of the digital modulation to create RF pulses. A digital-to-analog converter (28) converts the digital information to an analog RF pulse which is applied to a subject in an image region. The receivers each include an analog-to-digital converter (60) which digitizes the magnetic resonance signal emanating from the subject in the image region with four fold oversampling. A pair of FIR filters (62a, 62b) multiply the digital resonance signal by digital sine and cosine filter coefficients to create in-phase and out-of-phase digital magnetic resonance signal components. After additional digital filtering (64a, 64b), the digital in-phase and quadrature components are Fourier transformed (70) and accumulated in an image memory (72) to form an image representation.

Description: This invention relates to magnetic resonance systems. It finds particular application in conjunction with whole body magnetic resonance imaging systems and will be described with particular reference thereto. However, it is to be appreciated that the invention may also find application in conjunction with small bore imaging systems, magnetic resonance spectroscopy systems, and the like. Heretofore, medical magnetic resonance imaging systems have utilized analog transmitters and receivers. The transmitters commonly generated a radio frequency signal at the Larmor frequency of dipoles of interest in the imaging region. The radio frequency signal was shaped into pulses with selected frequency, phase, and amplitude characteristics and sent by way of a power amplifier to radio frequency transmitter coils. The pulses of frequency, phase, and amplitude modulation were effected with analog components. Frequency modulation was accomplished with a frequency synthesizer which generated an analog signal of selectable frequency. Although some frequency synthesizers had digital sections, the resultant frequency signal was analog. Pulse amplitude modulation was commonly conducted with double balanced mixers and analog gates or switches. Similarly phase modulation was achieved by phase shifting an intermediate radio frequency through the use of a phase splitter and combining it with an appropriate frequency to achieve the required Larmor frequency again through the use of a double balanced mixer. The use of these devices presented problems that double balanced mixers are inherently non-linear, e.g. square law devices, phase splitters had a wide error margin and analog gates tended to have leakage introducing unwanted characteristics in the transmitted radio frequency pulse. In order to operate the double balanced mixer in the most linear portion of its operating curve it was necessary to add external components to bias the input appropriately. Since the bias point could vary from unit to unit it was necessary to incorporate adjustable components. Similar methods were required to calibrate phase splitting devices and reduce leakage in analog gates. These adjustments not only increased the cost and complexity of the hardware and initial calibration, but also provided numerous unauthorized adjustment points. The analog magnetic resonance signals emanating from the subject were received and demodulated by an analog based receiver. Analog phase sensitive detectors produced sine and cosine related channels of analog signals. The sine and cosine analog signals were separately digitized and digitally processed to perform fast Fourier transform and other digital processing operations to create an image representation. Analog receivers were, again, non-linear and suffered the above described problems of non-linear components. Signals which bled through the analog components or devices became artifacts in the resultant image. The variations from device to device were more critical in the receiver. Failure to maintain channel to channel balance, amplitude, and phase consistency in the sine and cosine or real and imaginary channels also resulted in image artifacts. The analog receivers were further subject to DC level errors. Typically, twenty to thirty seconds were required before each scan in order to determine the actual, current DC level so that appropriate compensation could be made. EP-A-0411840 (Art. 54(3)EPC) discloses a receiver for a medical imaging apparatus which receives an input signal containing information in a band of frequencies contained within a bandwidth BW. The receiver includes means for demodulating the input signal to produce a resultant signal in which the information from the input signal lies within a band of frequencies centered at a frequency which is 1.5 times the bandwidth BW; and an analog to digital converter for sampling the resultant signal at a rate equal to twice the bandwidth BW and converting the samples into a digital signal. EP-A-0411840 also discloses a digital transmitter. An RF excitation pulse is generated by sequentially reading out a series of stored digital values, that represent the desired envelope. EP-A-0292064 discloses a magnetic resonance imaging device comprising a digital transmitter/receiver. The transmitter comprises a digital frequency synthesizer, a single sideband modulator and a selective power amplifier, the single sideband modulator being controlled by a phase locked loop oscillator. The amplitude of the transmitted signal is digitally controlled by multiplying the synthesized sinusoidal signal by a number which is stored in a register. The receiver comprises a selective preamplifier, a frequency mixing stage which is connected either to an output of the phase locked loop oscillator or to an output of the single sideband modulator, and an analog-to-digital converter which samples the output signal of the frequency mixing stage. The demodulation frequency for the receiver is chosen so that, after demodulation, a frequency band is obtained which is situated to one side of 0 Hz. According to the present invention there is provided a magnetic resonance imaging system comprising a clocking means (F) for generating clocking signals, a digital transmitter (B) which includes a digital modulation means for numerically modulating amplitude, phase, and frequency of a radio frequency signal that is transmitted into an examination region to excite and manipulate magnetic resonance, the digital modulation means being clocked by the clocking signals, the digital transmitter (B) further including a means for interrupting a receipt of clocking signals by the digital modulation means for creating pulses in the transmitted radio frequency signals, a digital receiver (C) for receiving magnetic resonance signals emanating from the examination region and converting the received signals to digital data and an image processing means (D) for processing the digital data into an image representation. One advantage of the present invention is that it improves the image quality and eliminates artifacts. Another advantage of the present invention is that it reduces component cost. Another advantage of the present invention is that it functions in real time without batch processing. Further advantages include simplified calibration and the elimination of service technician accessible adjustments. A magnetic resonance system in accordance with the present invention will now be described, by way of example, with reference to the accompanying drawings, in which:- Figures 1A and 1B together constitute a two part diagrammatic illustration of the magnetic resonance system; Figure 2 is a diagrammatic illustration of the system of Figures 1A and 1B modified for concurrently imaging multiple nuclei; and Figure 3 is a diagrammatic illustration of the system of Figures 1A and 1B modified for digital quadrature detection. Referring to Figures 1A and 1B, a magnetic field means A creates magnetic fields and magnetic field gradients through a subject in an examination region. A digital transmitter B receives digital frequency and phase control words, digitally processes the digital words, and transmits a corresponding radio frequency signal into the examination region. Analog radio frequency magnetic resonance signals emanating from the image region are digitized, demodulated, and divided into sine and cosine components digitally by a digital receiver C. Digital processing circuitry D digitally Fourier transforms and otherwise processes the digital magnetic resonance data from the receiver into an image representation. A sequence controller E sends the control words and other control signals to the digital transmitter, receiver, and processing circuitry to implement a selected one of a plurality of magnetic resonance imaging sequences. A common clock F synchronously clocks the sequence controller E, the transmitter B, the receiver C, and the image processing circuitry D. The magnetic field means A includes a plurality of resistance or superconducting magnets 10 which are controlled by a main magnetic field control means 12 to produce a generally uniform, static magnetic field through the examination region, commonly a 0.5, 1.0, or 1.5 Tesla field. A plurality of magnetic field gradient coils 14 are operated under the control of a gradient field coil controller 16 for selectively creating gradients across the main magnetic field. Preferably, gradients of selectable slope are selectively created along three mutually orthogonal axes, for convenience denoted as x, y, and z axes. The digital transmitter B includes a digital synthesizer chip 20 which has input registers for receiving (i) a digital frequency control or selection word or signal, (ii) a digital phase selection or control word signal, (iii) a digital offset or carrier frequency word or signal that identifies the carrier frequency, and (iv) a clock signal, e.g. 10 MHz. In the preferred embodiment, the synthesizer chip 20 is a numerically controlled modulated oscillator, NCMO. The output phase and frequency of the numerically controlled modulated oscillator is determined by the clock frequency, the frequency control word, the phase control word, and the offset frequency word received from the sequence controller E. The numerically controlled modulated oscillator output addresses a wave-form map stored in a programmable read only memory (PROM) 22 to translate sine wave phases to a corresponding amplitude. Optionally, the PROM may store a plurality of wave-forms, as may be appropriate to different imaging sequence one of which is selected by the sequence controller E. A digital modulating means 24 numerically modulates at least one of the amplitude, phase, and frequency of the transmitted radio frequency signal. The modulating means includes an amplitude modulation means 26 which multiplies the wave-form from the wave-form map by a digital amplitude profile word or signal from the sequence controller to form a pulse profile that digitally describes a selected pulse. In the preferred embodiment, the amplitude modulation means is a digital multiplier which multiplies each output point from the wave-form map by the digital values of the amplitude profile word. The multiplier 26 is clocked at the same speed as the numerically controlled modulated oscillator, such that each wave-form point is multiplied once by a corresponding amplitude profile value. Because the amplitude modulation is achieved numerically, the resultant digital pulse profile is highly linear and has substantially no carrier feed through. A first digital-to-analog converter 28 converts the digital pulse profile to a corresponding analog pulse profile. A clock signal interrupting means 30 selectively passes and blocks the clock signals from the digital modulating circuits to form pulses in the radio frequency signal. A mixer 32 mixes the analog transmitter pulse profile with an appropriate mixing frequency to produce the Larmor frequency for the selected nuclei. In the preferred embodiment, the selected nuclei are protons which have Larmor frequencies of 21.3, 42.6, and 63.9 MHz at magnetic fields of 0.5, 1.0, and 1.5 T, respectively. An analog band-pass filter 34 removes off frequency noise before the signal is amplified by a power amplifier 36. The amplified transmitter RF signal is applied to a radio frequency antenna 38 which broadcasts radio frequency energy of the corresponding phase, frequency, and amplitude into the examination region in order to induce magnetic resonance or manipulate resonating nuclei of the subject. On each cycle of the 10 MHz clock, data is transferred or pipelined between the stages. A latch 40 is clocked to receive and hold each digital wave-form word from the wave-form map to insert one clock cycle delay prior to converting from digital to analog by a digital-to-analog converter 42. This maintains a constant phase relationship between the two digital-to-analog converters. A mixer 44 mixes the analog waveform with a carrier frequency. This generates an analog reference signal of the same frequency and phase as the transmitter pulses. During the receive cycle, the base operating frequency of the NCMO 20 is lowered by 125 kHz. The analog magnetic resonance signal emanating from the subject in the examination region is picked up by the antenna 38 or a surface coil (not shown). The analog signal is amplified by an amplifier 50 and down converted to a frequency centered around 125 kHz by an analog mixer 52. A low-pass filter 54 eliminates frequency components that are outside of the bandwidth of the receiver. An analog-to-digital converter 60 digitizes the filtered NMR signal at a sampling rate or frequency selected in accordance with the down stream processing. The sampling theorem limits the highest carrier frequency that can be demodulated to one half of a fixed sampling frequency. The analog-to-digital converter is clocked to produce a four fold oversampling, i.e. four times the carrier rate. In the preferred embodiment, the sampling rate is 500 kHz which demodulates carrier frequencies into 125 kHz to produce a maximum bandwidth of 250 kHz. Demodulating the carrier is achieved by multiplying the received signal with in-phase (cosine) and out-of-phase (sine) components of the carrier rate. More specifically, the digital signal is conveyed to a pair of data signal processing chips 62a, 62b which operate as a finite impulse response (FIR) digital filter. The FIR filter performs a quadrature baseband demodulation and low-pass filtering in a single operation. Specifically, the FIR filter coefficients are multiplied by the sine and cosine of the demodulation frequency. Because the carrier is one fourth of the sampling frequency, the sine and cosine can be defined by four symmetric points, e.g. the 0, +1, 0, -1 points. This enables the FIR filter to use only rebinning, negation, and zero filling operations to multiply the sine and cosine. For a 125 kHz carrier and 500 kHz base sampling, the demodulated signal is formed from the digitized signal, S, as follows: Carrier Components Demodulated Signal Time (usec) In Phase Out of Phase Real Imaginary 2 +1 0 S O 4 0 +1 O S 6 -1 0 -S O 8 0 -1 O -S 10 +1 0 S O 12 0 +1 O S 14 -1 0 -S O 16 0 -1 O S . . . . . . . . . . . . . . After the demodulation operation the resultant data may be decimated to achieve varying NMR sampling rates. With a fixed base sampling rate, the NMR rate is restricted to integer multiples of this rate. In the above example, NMR sampling is restricted to 4, 6, 8, 10, 12, . . . µsec (or equivalently 250, 167, 125, 100, 83, . . . kHz). This gives signal in-phase data from one of the digital signal processing chips and quadrature or out-of-phase components from the other digital signal processing chip, both of which are centered around DC. In this manner, the equivalent of an analog phase sensitive detection is performed digitally. However, the digital operations are more precise over the entire imaging bandwidth and require no adjustment or DC offset correction. Because the signal was centered at 125 kHz, low frequency analog noise, which in an analog system would manifest itself as a DC artifact, is now outside of the bandwidth. Hence, no DC correction is required. A pair of digital filters 64a, 64b perform a second stage of digital filtering on the quadrature channels. The pair of digital filters use digital signal processing devices that provide variable sampling rates and function as the digital equivalent of a programmable analog anti-alias filter. More specifically to the preferred embodiment, each filter includes a second FIR filter each with 24 bit coefficients to operate at the selected data processing rate, e.g. 125 kHz. Each FIR filter rejects frequencies outside of the bandwidth set by the sampling rate with a digital equivalent of a low-pass filter and has the form: where: o(i) = the output signal value at time (i-1)*dt s(i) = the input signal value at time (i-1)*dt a(j) = the jth filter coefficient n = the total number of filter coefficients (also called "taps") dt = time between sampling points. The number of filter coefficients a(j) depends on the required filter properties: passband frequency, stopband frequency, baseband ripple, stopband attenuation. In general, as the passband is narrowed, the number of coefficients increases. The computational load need not increase because the multiplication/accumulations need only be done at the output, (i.e. the NMR sampling rate, rather than the input sampling rate). The variable sampling rate is chosen by decimation at the output of the filters. The bandwidth is inherently matched to the sampling rate. Different filter coefficients are loaded if the sampling rate or bandwidth is changed. A digital signal processor 66 provides the same coefficient a(j) and signal loading to the FIR filters 64a, 64b to assure that the same filtering is performed on each channel. The digital processing means D includes a digital fast Fourier transform means 70 which performs an on the fly transform of each sampled line of data or view. The Fourier transformed views are further processed, as is known in the art, and accumulated in an image memory 72 to form an image representation. The image representation in the image memory may be displayed on a video monitor 74 , stored on tape or disk, subject to further processing, or the like. The common clock F includes a high stability TTL oscillator, e.g. a 40 MHz oscillator 80. A clock generator 82 divides the frequency by two, four, and by 80 to produce the 20 MHz, 10 MHz, and 500 kHz clocking signals. With reference to FIGURE 2, a plurality of nuclei can be imaged concurrently. A plurality of digital transmitters B 1 , B 2 , ... B n are provided for exciting magnetic resonance in each of the plurality of nuclei. A corresponding plurality of digital receivers C 1 , C 2 , ... C n are provided for receiving and demodulating signals in accordance with the resonance frequency of each of the plurality of nuclei. By way of example, protons or hydrogen molecules have a resonance frequency at 63.9 MHz in a 1.5 T field and phosphorous has a resonance frequency of 26 MHz. The transmitter B 1 provides an output for exciting resonance and the protons and transmitter B 2 provides an output for exciting resonance in the phosphors. The two transmitters are operated under the control of the sequence controller E and the clock F such that each provides an output for exciting resonance in its corresponding nuclei simultaneously. The output resonance excitation signals are combined or summed and are amplified by the power amplifier 36 before being fed to the resonator 38. On the receiver side, the received resonance signal contains components at two Larmor frequencies. The signal is split in two and fed to receivers C 1 and C 2 for demodulation at the frequency corresponding to the phosphorous and hydrogen, respectively. The outputs of the two receivers are processed separately by the imaging processing circuitry D to generate separate digital representations of the baseband demodulated signals. The resultant images may be displayed on the video monitor 74 either side by side, superimposed, or the like. Rather than superimposing the resonance signals for each nuclei, the excitation and detection of each nuclei can be done in an interleaved manner in a single scan. The very high off-isolation of a digital transmitter enables the two transmitter outputs, i.e., the non-transmitting output of one and the resonance excitation output of the other, to be combined without any switching means for selecting among the transmitters. Second, the high accuracy and precision of the digital frequency generation enables the transmitters to be alternated with no, or substantially no, phase error. Such phase errors accumulate during the signal averaging process in a spectroscopic experiment. A digital transmitter's elimination of this error avoids unnecessary loss of the signal-to-noise ratio. Each of the receivers is again tuned to demodulate received signals at frequencies corresponding to a respective one of the excited nuclei. As yet another alternative, the imaging sequences for each nuclei can be conducted sequentially. With reference to FIGURE 3, a plurality of receivers can be connected with different antennae to receive different portions of the signal from common nuclei. The coils are positioned adjacent each other, e.g., sequentially along a patient's spine. In the illustrated embodiment, two coils are positioned coaxially to form a quadrature pair. Alternately, two planar coils may form the quadrature pair. Two receivers, C 1 and C 2 , are each connected with one of the two coils. The coefficients of the digital FIR filters are selected such that one is phase-shifted 90 degrees relative to the other. In this manner, the 90 degree phase shift is precisely conducted digitally rather than using passive components such as capacitors, inductors, or resistors on the coils. The digital signal combination and phase shifter is inherently more precise which leads to superior quadrature combination, hence an improved signal-to-noise ratio. Moreover, passive components are eliminated from the coils simplifying coil design and allowing high impedance circuits to be used more readily. The reduction in components on the coil also achieves a lower signal loss, less stray capacitance, and other undesirable real world phenomena.

Claims: A magnetic resonance imaging system comprising a clocking means (F) for generating clocking signals, a digital transmitter (B) which includes a digital modulation means (24) for numerically modulating at least one of the amplitude, phase, and frequency of a radio frequency signal that is transmitted into an examination region to excite and manipulate magnetic resonance, the digital modulation means (24) being clocked by the clocking signals, the digital transmitter (B) further including a means (30) for interrupting a receipt of clocking signals by the digital modulation means (24) for creating pulses in the transmitted radio frequency signals, a digital receiver (C) for receiving magnetic resonance signals emanating from the examination region and converting the received signals to digital data and an image processing means (D) for processing the digital data into an image representation. A system according to Claim 1, wherein the digital receiver (C) includes digital demodulating means (62a, 62b) for digitally quadrature baseband demodulating the digital data. A system according to Claim 1, wherein the digital receiver (C) includes combined digital demodulating and filtering means (62a, 62b) for digital quadrature demodulating and low-pass filtering the digital data. A system according to Claim 1, wherein the digital receiver (C) includes finite impulse response digital filter means (62a, 62b) for digitally multiplying the digital data with in-phase and out-of-phase components of a carrier to demodulate and filter the digital data. A system according to Claim 4, wherein the digital data is sampled at four times the carrier and the in-phase and out-of-phase components are sine and cosine components which take only -1, 0, and +1 values, the digital data demodulation being performed by rebinning, negation, and zero-filling. A system according to Claim 4 or Claim 5, wherein the digital receiver (C) includes sampling means (60) for digitizing the received analog resonance signal at a selected sampling rate and a digital filter (64a, 64b) having a filter cut-off frequency that matches the sampling rate. A system according to Claim 1, wherein the digital receiver (C) includes analog-to-digital convertor means (60) for converting analog resonance signals to digital resonance signals and digital signal processing means (62a, 62b) for digitally multiplying the digital resonance signals by sine and cosine filter coefficients, respectively to generate in-phase and quadrature digital magnetic resonance signal components. A system according to Claim 7, wherein the analog-to-digital converter means (60) performs a four fold sampling and the digital signal processing means (62a, 62b) includes a finite impulse response means (62a, 62b) that negates, zeroes, and rebins the digital magnetic resonance signals for effectively multiplying it by +1, 0, and -1. A system according to any one of Claims 1 to 8, wherein the digital transmitter (B) includes multiplying means (26) for digitally multiplying a first digital signal indicative of a wave-form characteristic by a digital signal indicative of a selected modulation. A system according to any one of Claims 1 to 8, wherein the digital transmitter (B) includes a numerically controlled modulated oscillator (20) which receives digital frequency and phase control signals and a clocking signal. A system according to Claim 10 further including a wave-form memory means (22) which is addressed by the output of the numerically controlled modulated oscillator (20) to translate the digital input data to a waveform of appropriate phase and frequency. A system according to Claim 11 further including multiplying means (26) for multiplying the waveform digitally by a digital amplitude profile such that the resultant waveform is digitally amplitude modulated. A system according to Claim 12 further including a mixer (32) for combining the amplitude modulated waveform with a carrier frequency and a power amplifier (36) for amplifying the mixed signal and conveying it to the radio frequency antenna (38). A system according to Claim 1, wherein the clock generator (F) is a common clock generator (F) which generates clock signals for synchronously clocking the digital transmitter (B or B 1 ), the digital receiver (C or C 1 ), and the digital image processing means (D). A system according to Claim 14 further including at least one additional digital receiver (C 2 - C n ), which is clocked by the common clock generator (F). A system according to Claim 15, wherein the digital receivers (C 1 - C n ) are connected with different coils (38 1 , 38 2 ) and wherein each receiver (C 1 - C n ) includes a finite impulse response digital filter means (62a, 62b) for digitally multiplying the digital data with in-phase and out-phase components of a carrier to demodulate and filter the digital data, the finite impulse response digital filter means (62a, 62b) of each receiver (C 1 - C n ) having different coefficients which have a 90 degree phase shift therebetween. A system according to Claim 15 further including at least one additional transmitter (B 2 - B n ) which is clocked by the common clock generator (F), the digital transmitters (B 1 - B n ) each exciting different nuclei to resonance and the digital receivers (C 1 - C n ) each receiving and demodulating signals from one of the nuclei.

Inventors: BLAKELEY DOUGLAS M, DENISON KENNETH S, FLUGAN DAVID C, HOLLAND G NEIL, STAUBER JOHN R, BLAKELEY, DOUGLAS M., DENISON, KENNETH S., FLUGAN, DAVID C., HOLLAND, G. NEIL, STAUBER, JOHN R.
Applicants: PICKER INT INC, PICKER INTERNATIONAL, INC.
Application year: 1990
Main classification: G01R33
IPC codes: G01R33, A61B5