A TE01 mode dielectric resonator circuit

Abstract: A resonator element (5) formed of a half or a quarter of dielectric cylinder contacts an electrically conductive plane (11) via the resonator element's radially cut side (5′) which includes the axis of the cylinder, accordingly, resonates in TE 01δ -mode. On an opposite side of the electrically conductive plane there is provided an unbalanced transmission line, for example, of a strip line type or a coaxial line type. An end of the transmission line is electromagnetically coupled via a dielectric material in the transmission line or directly with the radially cut side (5′) of the resonator element (5) through an opening provided on the electrically conductive plane (11). Coupling circuit according to the present invention allows a compact overall circuit configuration.

Description: BACKGROUND OF THE INVENTION Field of the Invention The present invention relates to a dielectric resonator according to the preamble of claim 1 which is known from GB-A-2 201 045 or from the US-A-4 423 397. Description of the Related Art A prior art TE 01δ mode dielectric resonator employed in a bandpass filter and the method of coupling with its external circuit are shown in FIG. 1 through FIG. 3. In FIG. 1, between two standard waveguides (i.e. TE₁₀ mode waveguides) 1 and 1′ there is connected a second waveguide 2 which is in a cut-off state for the electromagnetic wave to be now transmitted through the standard waveguides 1 and 1′. A TE 01δ mode cylindrical dielectric resonator element 3 is installed in the second waveguide 2 via a metal stage 4 mounted on its side wall parallel to the larger side walls of the standard waveguides 1 and 1′. The resonator element 3 is coupled magnetically, i.e. via magnetic flux, with both the standard waveguides 1 and 1′, so as to allow only the resonator element's resonant frequency to transmit through the cut-off waveguide 2. In this circuit configuration, the stage 4 causes an increase in space occupancy of the circuit. In order to reduce the space occupancy, a configuration shown in FIG. 2 has been proposed, such as disclosed in Japanese TokuKai Hei-1-144701. In this circuit configuration, a half-cut cylindrical dielectric resonator element 5 has its flat surface adhered to a shorter side wall of the cut-off waveguide 2, and is magnetically coupled with the standard waveguides 1 and 1′. In FIG. 3, a half-cut dielectric resonator element 5 is adhered on an inner wall of a metal case 7 so as to interconnect coaxial lines 6 and 6′. In this circuit configuration, an extension of each of the inner conductors of the coaxial lines 6 and 6′ is terminated on the metal case 7 and forms a loop 6a which is magnetically coupled with the half-cut cylindrical resonator element 5. However, there are problems in that in the FIG. 2 configuration the overall circuit size is little reduced even though the resonator element is reduced into a half size; and in the FIG. 3 configuration the loops 6a require the space in the case 7. The same problem is in a circuit configuration employing a quarter cut TE 01δ -mode dielectric resonator element reported in "IEEE Transaction on Microwave Theory and Techniques", vol. MTT-35, No. 12, Dec. 1987, p.1150-1155. Thus, there is no much likelihood of further size reduction in the above-described circuit configuration. Therefore, a new coupling circuit which can enjoy the advantage of the compact half or quarter cut cylindrical dielectric resonator has been expected. SUMMARY OF THE INVENTION It is a general object of the invention, therefore to provide a compact circuit configuration for coupling a half or quarter-cut cylindrical TE₀ 1δ -mode dielectric resonator to an outer transmission line. It is another object of the invention to provide a circuit configuration suitable for mounting a half or quarter-cut cylindrical TE 01δ -mode dielectric resonator onto a printed circuit board. This object is solved by a dielectric resonator of the initially defined type having the characterizing features of claim 1. Preferred embodiments are listed in the dependent claims. The above-mentioned features and advantages of the present invention, together with other objects and advantages, which will become apparent, will be more fully described hereinafter, with reference being made to the accompanying drawings which form a part hereof, wherein like numerals refer to like parts throughout. BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 schematically illustrates a prior art bandpass filter employing a TE 01δ -mode cylindrical resonator element, where the side-walls of the waveguides are not shown for simplifying the drawing; FIG. 2 schematically illustrates a prior art bandpass filter employing a TE 01δ -mode half-cut cylinder resonator element, where the side-walls of the waveguides are not shown for simplifying the drawing; FIG. 3 schematically illustrates a prior art bandpass filter employing a TE 01δ -mode half-cut cylindrical resonator element, connected with coaxial transmission lines; FIG. 4 schematically illustrates a first preferred embodiment of the present invention employed for connection with coaxial transmission lines; FIG. 5 schematically illustrates a second preferred embodiment; FIG. 6 shows a vertically cut side view of a third preferred embodiment of the present invention; FIG. 7 shows an inner side plan view of a ceramic substrate employed in FIG. 6 embodiment; FIG. 8 shows a perspective view of the components employed in FIG. 6 embodiment; FIG. 9 shows an outer side plan view of the ceramic substrate employed in FIG. 6 embodiment; FIG. 10 shows a perspective view of the complete FIG. 6 filter; FIG. 11 shows bandpass characteristics of FIG. 6 filter; FIG. 12 shows an enlargement of FIG. 11 bandpass characteristics in the vicinity of the resonant frequency; FIGs. 13(a) and 13(b) show a fourth preferred embodiment of the present invention; FIG. 13(c) show the opposite side of the ceramic substrate shown in FIG. 13(b); FIG. 14 shows a fifth preferred embodiment of the present invention; and FIG. 15 shows a sixth preferred embodiment of the present invention. DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 4(a) shows a cross-sectional plan view, and FIG. 4(b) shows a cross-sectional side view, of a first preferred embodiment of the present invention. A dielectric resonator element 5 is formed of a dielectric material, such as (ZrSn)TiO₄ whose dielectric constant is as high as 36.5 or Ba₂Ti₉O₂₀ whose dielectric constnat is 39.8. The dielectric resonator 5 is in the shaped of a half-cut cylinder having a flat side 5′ which includes the axis (not shown in the figure) of a dielectric cylinder of, for example, 6 mm diameter. The flat side 5′ is referred to hereinafter as a radially cut side. The half-cut cylinder is also cut with two planes orthogonal to the axis of the cylinder so as to leave, for example, 2.3 mm thickness. The radially cut side 5′ is adhered to a metal wall 11 of a resonator base 12 typically with a generally available epoxy resin. The metal wall 11, being electrically conductive, acts as a mirror to form an image of the half-cut cylinder dielectric resonator element 5, so that the half-cut cylindrical dielectric resonator element 5 resonates in a TE 01δ -mode like a fully cylindrical dielectric resonator element. Resonant frequency of the resonator element varies depending on the element's dimensions and the dielectric constant of the element's material. First and second coaxial transmission lines 14 and 15, each having typically 50 ohm characteristic impedance, are provided vertically to the metal wall 11 through the resonator base 12. Each of coaxial transmission lines 14 and 15 is typically composed of 2.1 mm outer diameter, 0.63 mm inner conductor diameter, and Teflon (CF₄) filled therebetween. End 16 and 17 of each inner conductor 14′ and 15′ of respective coaxial transmission lines 14 and 15 faces the radially cut side 5′ via a predetermined distance d (denoted in FIG. 4(b)), for example, 0.5 mm. An electromagnetic wave signal transmitted on the inner conductor 14′ of the first coaxial transmission line 14 is electromagnetically coupled to the radially cut side 5′ of the resonator element 5 via capacitance formed at the above-described distance. That is, current flowing from the inner conductor 14′ through the capacitance excites the resonator element 5, and further flows along the TE 01δ mode electric field 8 in the resonator element 5 shown in FIG. 4(a). The term "coupling" is referred to so as to express this phenomena. This current reaches the inner conductor 15′ of the second coaxial line 15, in the same but reverse way as the first coaxial line 14, only when the frequency of the signal causes TE 01δ mode resonance in the resonator element 5. Other frequency than the resonant frequency does not reach the second coaxial line 15 and reflects back to the first coaxial line 14. Thus, the resonator element 5 acts as a band pass filter. The other ends of the coaxial lines 14 and 15 are connected to coaxial connectors 17 and 18, respectively. Thus, the circuit of FIG. 4 can be handled as an independent filter, easily detachable from coaxial cables. Metal cap 13 is electrically connected, for example soldered, to the resonator base 12 so that the resonator element 5 is confined in its cavity as well as shielded from other circuits. Electric field strength expressed with density of electric fields 8 is weak at the peripheral portion or at the centre portion of the half-cut cylinder 5. A coaxial transmission line connected to the higher electric field portion provides a closer coupling, as well as less coupling at a weaker electric field portion. Therefore, the coupling between the transmission line and the resonator element 5 can be varied by choosing the location of the transmission lines 14 and 15 along the radial direction of the dielectric cylinder. The coupling between the transmission line and the resonator element 5 can be adjusted also by the capacitance value at the distance between the inner conductor ends 16 or 17 and the radially cut side 5′ of the resonator element 5. The closer coupling between the transmission line and the resonator element 5 provides the wider pass-band width of the filter. In order to achieve impedance matching of the input transmission line 14, locations of the two transmission lines 14 and 15 are preferably chosen at the symmetric positions with respect to the axis of the resonator element 5. FIG. 5 shows a second preferred embodiment of the present invention, as a modification of FIG. 4 first preferred embodiment. Each of inner conductors 14′ and 15′ and their ends 16′ and 17′, of the coaxial lines, are printed on a ceramic substrate (not shown in the figure). The ends 16′ and 17′ are made wider than the 50 ohm transmission line portion 14 and 15 so as to form a properly increased capacitance with the radially cut side 5′ of the resonator element 5. In order to adjust the capacitance, the shape of the ends 16′ and 17′ can be adjusted by removing the printed conductor by means of, for example, sand blasting. Advantage of FIG. 5 configuration is in that the coupling capacitance value can be precisely controlled. A third preferred embodiment of the present invention, where the input and output transmission line circuits are formed of strip line type transmission lines, is schematically illustrated in FIG. 6 showing a vertically cut cross-sectional view; FIG. 7 showing an inner surface plan view of its ceramic substrate; FIG. 8 showing a perspective view of the composing elements; FIG. 9 showing an outer surface plan view of the ceramic substrate; and FIG. 10 showing a perspective view of the complete filter mounted on a mother board. According to a widely employed method, electrically conductive planes 22a of, for example, copper, is formed upon a surface of, for example, a 0.65 mm thick alumina ceramic substrate 22, and is provided with two openings 22h of typically 0.8 mm diameter and spanned by 2 mm, by chemical etching or sandblasting so as to expose part of the ceramic substrate 22, while circular patterns 22b and 22c, as coupling electrodes, are left at the centre of each opening. In the same way, on the other surface of ceramic substrate 22, there are formed an input strip electrode 22f, an output strip electrode 22g, each having 0.6 mm width, and a ground plane 22a′. Shorter sides of substrate 22 may be also coated with an electrically conductive material so that both the ground planes 22a and 22a′ are electrically connected. Each of strip electrodes 22f and 22g, together with this side of ground plane 22a and the 0.65 mm thick ceramic substrate therebetween, constitute strip-line type 50 ohm transmission line. Hatched portions in FIGs. 4 and 5 indicate the exposed ceramic substrate 22. At the centers of coupling electrodes 22b and 22c, there are provided through-holes 22d and 22e coated with electrically conductive material so as to electrically connect each of the coupling electrode 22b and 22c to ends of the strip electrodes 22f and 22g, respectively. Each of the opposite ends 22f′ and 22g′ of strip electrodes 22f and 22g vertically extends along thin side of the ceramic substrate 22 so as to be terminals to be connected with external circuit by soldering. Resonator element 21a is substantially the same as the resonator element 5 used in the first preferred embodiment. The radially cut side 21a-1 of the resonator element 21a is adhered onto the metal plane 22a as well as the openings 22h, in the same way as those of FIGs. 4 and 5. A metal cap 23 is soldered onto the metal plane 22a in order to shield the resonator element 21a from the other circuits, as denoted with the numeral 24. Thus completed filter unit 21 is mounted onto a mother circuit board 28 by soldering the ground planes 22a and 22a′ onto a ground plane 29, as well as terminals 22g′ and 22f′ to a strip electrode 26, each of a mother circuit board 28. Degree of the coupling between the transmission line and the resonator element is determined by the size of openings 22h, the size of the coupling electrodes 22b and 22c and the location of the openings measured from the axis of the half cylinder. The coupling electrodes 22b and 22c provide relatively large capacitance value, resulting in a close coupling with the resonator element 21a. In order to achieve relatively loose coupling with the resonator element 21a, the coupling electrodes 22b and 22c and the through-holes 22d and 22e may be omitted. This case is not shown in the figure. In this case, the degree of the coupling is determined by the capacitance between the strip electrode and the resonator element, that is, by the size of the opening, the area of the strip electrode facing the resonator electrode through the opening, and the thickness as well as dielectric constant of the ceramic substrate 22 existing therebetween. Bandpass characteristics of FIG. 6 filter are shown in FIGs. 11 and 12. FIG. 11 shows frequency characteristics from 1 to 26 GHz, where a peak at 9.848 GHz is of the TE 01δ mode resonance of the resonator element, while other peaks existing at higher frequency band than the TE 01δ mode resonance are of higher mode resonances of the resonator element and of the resonance of the cavity formed with cap 23. FIG. 12 shows an enlargement of the FIG. 11 bandpass characteristics in the vicinity of the TE 01δ mode resonance. The - 3 db band width is 12.8 GHz for the centre frequency 9848.425 MHz, and the insertion loss is 16.5 db. The insertion loss will be much reduced by employing more suitable material for adhering the resonator element to the substrate. Size of bandpass filter unit 21 shown in FIG. 6, used for 10 GHz band, achieved 7 mm high x 8 x 14 mm cap and 12 x 18 mm substrate. Thus, the filter volume is as small as approximately 1.4 cc, which is a half of 2.8 cc of case 7 in FIG. 3 of the prior art filter employing coupling loops. Moreover, FIG. 6 structure is suitable for being easily handled and mounted on a strip line type mother circuit board, which is the most commonly employed today, as well as allows the mother board to be compactly finished. A variation of the substrate embodied in the third preferred embodiment is shown in FIGs. 13(a) and 13(b). FIG. 13(b) explains assembling of the components. FIG. 13(c) shows the opposite surface of ceramic substrate 32 shown in FIG. 13(b). Cap 23 and resonator element 21a are substantially the same as those of FIG. 6. Ground planes 32a and 32a′ coated on the both surfaces of ceramic substrate 32 are electrically connected with each other via a plurality of through-holes 37 provided through the ceramic substrate 32 or via metal coat on the short sides of the ceramic substrate 32, and are soldered to a metal substrate 31. Metal substrate 31 is provided with two channels 43, which are, for example, 3 mm wide, 0.7 mm deep, and extend so as to face the strip electrodes 34. Between the two channels there is left a 1 mm wide bank 36. When ceramic substrate 32 is fixed onto metal substrate 31, the strip electrodes 34 are electromagnetically shielded in channels 33, respectively. Bank 36 act as an electromagnetic shield between input and output transmission lines 34. Strip electrodes 34 do not need extended portion 22f′ and 22g′ along the short sides of the ceramic substrate 22 as in FIG. 8. However, each end of strip electrodes 34 is extended with ribbon electrode 35 soldered thereto. Metal substrate 31 having the filter unit 30 thereon is fixed to a mother board (not shown in the figure) with screws 38 penetrating the openings provided on the metal substrate 31, then the ribbon electrodes 35 being flexible are easily soldered to a circuit on the mother board. This configuration allows an easy handling as well as quick mounting of the filter unit onto the mother board. A fourth preferred embodiment of the present invention is shown in FIGs. 14, where a plurality of the resonator elements 43A through 43C are employed in a single case 412. FIG. 14(a) shows a perspective view of the filter unit, whose top lid 412′ is disassembled. FIG. 14(b) shows a cross-sectional plan view of FIG. 14(a) filter. Each of the resonator elements 43A through 43C is essentially the same as that of FIG. 4 first preferred embodiment. Radially cut sides 42A, 42B and 42C of respective resonator elements 43A through 43C are adhered in line onto a metal wall 41 of case 412. A coaxial input terminal 417 according to the structure of FIG. 4 first preferred embodiment or FIG. 5 second preferred embodiment is arranged so as to couple the first resonator element 43A, at a farther side than the axis of the half cylinder of the resonator element 43A from the next resonator element 43B. The resonator element 43B located between the first and the last resonator elements is provided with no external coupling means through the wall 41. Each of the resonator elements 43A through 43C is mutually coupled with the adjacent resonator element by magnetic flux 49A and 49B of the TE 01δ mode as shown with dotted lines. Signal input from the input terminal 417 exciting the first resonator element 43A thus propagates along on each resonator element to the last resonator element 43C. A coaxial output terminal 418 similar to the input terminal 417 is provided so as to couple the last resonator element 43C, at the farther side from the previous resonator element 43B with respect to the axis of the half cylinder of the resonator element 43C. Thus, only the resonant frequency of the resonator elements 43A through 43C can be output from the output terminal 418. Degree of the mutual coupling between the neighbouring resonator elements determined by their distance determines the filter's pass-band width. A metal lid 412′ covers the top opening of the case 412. Metal screws 419A through 419C are provided in screw holes on metal lid 412′, and extends therefrom to over respective resonator elements. Resonant frequency of each resonator element can be finely adjusted by rotating the corresponding screw. The FIGs. 14 configuration is advantageous in that the space occupied by the coupling loops from/to the input/output circuit can be saved. It is apparent that FIG. 6 strip-line type input/output circuit can be also embodied in FIG. 13 multiple resonator element configuration, though no figure is given therefor. Though in FIGs. 14 fourth preferred embodiment the input and output terminals 417 and 418 are located respectively farther sides than each element axis, it is apparent that the input and/or output terminal(s) may be located nearer side than respective element axis as denoted with arrows 417′ and 418′. FIG. 15 shows a filter unit as a fifth preferred embodiment of the present invention. This configuration is suitable for a use in relatively low frequency band, such as below several hundreds Mega-Hertz band. Therefore, sizes of resonator element 50, ceramic substrate 51 and cap 52 are larger than those of FIG. 4 or FIG. 6 configuration; however the structures are quite similar thereto, except that the outer surface 51′ of substrate 51 has no coaxial lines nor strip electrodes. Electrically conductive through-holes 53 are provided through the ceramic substrate 51 so as to face the centers of the openings of the metal plane (not shown in the figure) on the inner surface 51″ of the substrate. Diameter of the through-holes, locations of the through-holes, and the distance between the ends of the through-holes and the radially cut side of the resonator, determine the degree of the coupling. Therefore, coupling electrodes may be additionally provided at the ends of the through holes as the FIG. 7 configuration. Electrically conductive leads 54 are soldered to the through-holes 53, as input and output terminals of the filter unit from and to other circuit. When a loose coupling is required, the above-described electrically conductive through-holes may be omitted, and a coupling electrode (not shown in the figures) may be provided on the outer surface 51′ of the ceramic substrate 51 in place of the through-holes. Then, leads 54 are soldered to the coupling electrodes on the outer surface 51′. Outer ground plane (not shown in the figure) coated on the outer surface 51′ of the substrate 51 is connected to inner ground plane via the electrically conductive through-holes (not shown in the figure) provided through ceramic substrate 51 or via metal coating (not shown in the figure) on the short side of the ceramic substrate 51. A grounding lead 55 is soldered to the outer ground plane at the centre of input/output leads 54. The grounding lead 55 located between input and output leads 54 is effective to electromagnetically shield the two leads 54. The grounding through-holes may be omitted, when the inner ground plane is grounded by other means. Grounding lead 55 may be omitted, when the ground plane 51″ can be grounded by other means. In addition to the advantage of the filter's less space occupancy, less number of the components is advantageous for cost reduction of the filter. Though a half-cut cylinder type resonator element is referred to in the above preferred embodiments, it is apparent that the concept of the present invention can be embodied for coupling the input/output circuit to a quarter-cut cylinder resonator element. The quarter-cut cylinder resonator element is such that two of the radially cut sides, each including the axis of the cylinder and orthogonal to each other, cut a dielectric cylinder so as to leave a quarter of the cylinder. The radially cut sides are contacted respectively with two metal walls orthogonal with each other. Each metal wall acts as mirror to form an image of the quarter cylinder so that the quarter-cut cylinder resonates equivalently in the TE 01δ mode of a complete cylinder. Quarter-cut cylinder resonator elements are reported in the above-cited IEEE Transaction. When a quarter-cut cylinder resonator element is provided with both the input and output terminals, the terminal is provided on each of the two orthogonally arranged metal walls. Though in the above-described preferred embodiments a radially cut side of the resonator element is contacted with a metal wall, it is apparent that radially cut side of the resonator element may be metalized with an electrically conductive material, excepting the openings for the electrostatic coupling. The metalization is carried out by a generally employed technique, such as plating, sputtering, sintering or printing of copper, gold or silver, etc. The metalized side of the resonator element may be further contacted with the metal wall referred to in the above embodiments, or may be directly employed for constituting the transmission line. The metalization of the resonator element reduces improves the insertion loss in the bandpass characteristics caused from the used of organic adhesive material. The many features and advantages of the invention are apparent from the detailed specification and thus, it is intended by the appended claims to cover all such features and advantages of the system which fall within the scope of the invention. Further, since numerous modifications and changes may readily occur to those skilled in the art, it is not desired to limit the invention to the exact construction and operation shown and described, and accordingly, all suitable modifications and equivalents may be resorted to, falling within the scope of the invention.

Claims: A dielectric resonator comprising: a resonator element (5, 21a, 50, 43) formed of a part of a dielectric cylinder, said dielectric cylinder being cut with a radially cut side (5′, 42, 21a-1) including the axis of said cylinder, said radially cut cylinder being further cut with two planes orthogonal to said axis; an electrically conductive plane (11, 22a, 32a, 41), a first surface of which contacts said radially cut side (5′, 21a-1, 42) of said resonator element (5, 21a, 43, 50), whereby said resonator element resonating with a radio frequency signal equivalently in TE 01δ -mode, said electrically conductive plane (11, 22a, 32a, 41) having an opening (22h); and a transmission line (14, 15, 22f, 22g, 34, 54, 417, 418) located opposite from said resonator element (5, 21a, 43, 50) with respect to said electrically conductive plane (11, 22a, 32a, 41), said transmission line being operatively connected to said opening, whereby an electromagnetic wave (8) carried on said transmission line is coupled via said opening to said resonator element; characterized in that said radially cut side (5′, 42, 21a-1) of said resonator element faces said opening. A dielectric resonator recited in claim 1, wherein said resonator element (5, 21a, 43, 50) is formed of a radially-cut half of said dielectric cylinder. A dielectric resonator recited in claim 1, wherein said resonator element is formed of a radially cut quarter of said dielectric cylinder, comprising two of said radially cut sides orthogonal with each other, said radially cut sides contacting first surfaces of two of said electrically conductive planes respectively. A dielectric resonator recited in claim 1, wherein said electically conductive plane (11, 22a, 32a, 41) is a metal plate supporting said resonator element (5, 21a, 43, 50). A dielectric resonator recited in claim 4, wherein said radially cut side (5′, 42, 21a-1) of said resonator element (5, 21a, 50, 43) is adhered to said metal plate. A dielectric resonator recited in claim 1, wherein said electrically conductive plane (11, 22a, 32a, 41) is formed of a metal film plated on said radially cut side (5′, 42, 21a-1). A dielectric resonator recited in claim 1, wherein said electrically conductive plane (11, 22a, 32a, 41) is formed of a metal deposition sputtered on said radially cut side (5′, 42, 21a-1). A dielectric resonator recited in claim 1, wherein said electrically conductive plane (11, 22a, 32a, 41) is formed of metal powder painted on said radially cut side (5′, 42, 21a-1). A dielectric resonator recited in claim 1, wherein said electrically conductive plane (11, 22a, 32a, 41) is metal film sintered on said radially cut side (5′, 42, 21a-1). A dielectric resonator recited in claim 1, wherein said transmission line (14, 15, 22f, 22g, 34, 54, 417, 418) is an unbalanced transmission line. A dielectric resonator recited in claim 10, wherein said unbalanced transmission line is a strip line type transmission line (22b, 22c), said strip line type transmission line being formed with a strip electrode (22f, 22g) and a second surface, opposite to said first surface (22a), of said electrically conductive plane and a dielectric layer (22) between said strip electrode (22f, 22g) and the second surface of said electrically conductive plane, said resonator element being electromagnetically coupled with an end (22b, 22c) of said strip electrode via said dielectric layer (22). A dielectric resonator recited in claim 11, wherein said end of said strip electrode extends through said dielectric layer (22) towards said opening (22h). A dielectric resonator recited in claim 10, wherein said unbalanced transmission line is a coaxial line (14, 15), an outer conductor (15) of said coaxial line being electromagnetically connected to said electrically conductive plane (11), an inner conductor (14′, 14′) of said coaxial line (14, 15) being electromagnetically coupled to said resonator element (5) via said opening. A dielectric resonator recited in claim 1, wherein a second one of said opening is additionally provided on said electrically concuctive plane. A dielectric resonator recited in claim 14, wherein said openings are located at essentially same distance from said axis of said cylinder. A dielectric resonator recited in claim 1, wherein said resonator further comprising a cap (13, 23, 52) containing said resonator element (5, 21a, 50), formed of an electrically conductive material and electrically connected to said electrically conductive plane (11, 22a, 32a).

Inventors: ASHIDA HIDEO, KONDO YASUYUKI, SOGO HIROYUKI, SUGAWARA HIDEO, ASHIDA, HIDEO, KONDO, YASUYUKI, SOGO, HIROYUKI, SUGAWARA, HIDEO
Applicants: FUJITSU LTD, FUJITSU LIMITED
Application year: 1990
Main classification: H01P7
IPC codes: H01P7