Direct-conversion FSK receiver having a DC output independent of frequency drift

Abstract: A receiver for FSK signals comprises means (10 to 22) for receiving the FSK signals and producing quadrature related first and second frequency down-converted difference signals, means (24,25) for amplitude limiting each of the first and second difference signals to form respectively first and second square wave signals, means (26,27,28,30) for sampling the first and second square wave signals at changes in polarity of the second and first square wave signals, respectively, means (32) for combining said samples and memory means (34), for example a hysteresis circuit (Figure 3 - not shown), responsive to a change in polarity in the combined signal for producing a first substantially constant dc output until the next following change in polarity is detected whereupon a second substantially constant dc output is produced. By using memory means (32,34) rather than a filter as is used in the prior art, reactive components are avoided making it less expensive to integrate and also a fixed level bit slicer can be used.

Description: The present invention relates to radio receiver for FSK signals, such a receiver may be used as a paging receiver. British Patent Specification 2,143,386A discloses a radio receiver suitable for use with FSK or angle modulated signals. The radio receiver comprises a frequency down-conversion stage which utilises direct conversion techniques to provide zero IF quadrature baseband signals I and Q. The I and Q signals are limited in respective limiting amplifiers to form corresponding I and Q square wave signals. Pulse forming networks, for example differentiators, are coupled to the outputs of the respective limiting amplifiers to derive respective first and second pulse sequences, the outputs of which represent the polarity changes in the corresponding I and Q square wave signals. First and second multipliers are provided in which the I and Q square wave signals are multiplied by the second and first pulse sequences, respectively, to produce two pulse trains. Means are provided for combining the two pulse trains to give a digital output signal the frequency of which is equal to the sum of the frequencies of the two pulse trains. The digital output signal is applied to a low pass filter which produces a simulation of the original baseband modulation. More particularly the pulses in the digital output signal are smoothed to give approximately a dc level, the magnitude of which depends on the pulse frequency which in turn depends on the frequency of the beat note, that is the frequency difference between the receiver's local oscillator frequency and the frequency of the modulated signal. The described receiver in principle functions satisfactorily if there is no drift between the local oscillator and modulated signal frequencies. However, in a practical situation drift will occur and the effect will be that the beat frequencies corresponding to binary "1" and "0" will be different. As a consequence of the low pass filtering the magnitude of the dc level will vary in accordance with the drift. In order to detect the value of the modulated signal a dynamic bit slicer is required. Dynamic bit slicers are complicated to design and consume a relative large current which undesirably increases the battery drain. The turn-on time of the receiver is lengthened by the use of the dynamic bit slicer which is a disadvantage when the receiver is operating in a battery economising regime. A reason for using a zero IF receiver is that it can be largely fabricated as an integrated circuit. However, as is well known, reactive components such as capacitors are expensive to integrate and if possible are avoided. Using a low pass filter to obtain an output dc signal mitigates against this as it requires one or more capacitors. GB-A-2057820 discloses a radio receiver for FSK signals which comprises means for producing quadrature related, amplitude limited zero IF signals which are applied to a digital demodulator comprising D-type flip-flops, NOR gates and set/reset flip-flops. The manner of connection of the D-type flip-flops and NOR gates is such that there is an inherent delay before the change in polarity of the output occurs as a result of a change in the input. An object of the present invention is to overcome these disadvantages in the design of a radio receiver. According to the present invention there is provided a receiver for FSK signals, comprising means for receiving the FSK signals and producing quadrature related frequency down-converted signals, means for amplitude limiting each of the frequency down-converted signals to form first and second square wave signals and demodulating means for producing an output signal, characterised in that the demodulating means comprises means for sampling the first and second square wave signals at changes in polarity of the second and first square wave signals, respectively, and for producing respective sequences of pulses having a duration less than the period of said first or second square wave signal, means for combining the sequences of pulses to provide an output which changes polarity at the occurrence of the first pulse from the sampling means following a change of frequency in the received FSK signal and means for providing an analog output representative of the transmitted base band signal, said last means comprising memory means responsive to the occurrence of a change in polarity in the combined signal for producing a first substantially constant dc output until the next following change in polarity is detected, and responsive at the detection of said next following change in polarity for producing a second substantially constant dc output. By the sampling means producing sequences of pulses having a duration less than the period of said first or second square wave signal and the output of the combining means changing polarity at the occurrence of the first pulse from the sampling means following a change in frequency of the received FSK signal, there is virtually no delay in detecting a change of frequency in the received FSK signal. Further by providing memory means responsive to a polarity change, which memory means may comprise a hysteresis circuit, the output signal voltage corresponds to the amplitude limited voltage indicative of a binary "1" or "0" and remains at that value until there is a polarity change. As a result the output signal voltage is, within the tolerances specified, independent of frequency drift. This means that a fixed level bit slicer can be used to detect the frequency modulation. A fixed level bit slicer is relatively simple to design and is economical in its current consumption. The memory means does not have either a frequency or impulse response and consequently does not require separate reactive components, such as capacitors, and is therefore cheaper to integrate. The means for sampling may include means for generating pulses having a duration of substantially one hundredth of the period of the first or second square wave signal. In one embodiment of the present invention the means for sampling comprises first means for multiplying the second square wave signal by pulses generated in response to changes in polarity in both directions of the first square wave signal to produce a first sequence of pulses and second means for multiplying the first square wave signal by pulses generated in response to changes in polarity in both directions of the second square wave signal to produce a second sequence of pulses. In another embodiment of the present invention, means are provided for providing third and fourth square wave signals which are inverted versions of the first and second square wave signals, respectively, in that the means for sampling comprises means for sampling the occurrence of alternate edges having the same polarity change in the first, second, third and fourth square wave signals and producing respective first to fourth sequences of pulses and a plurality of logic gates operative on the first to fourth square wave signals in response to said first to fourth sequences of pulses to provide further sequences of pulses which are applied to said combining means. The logic gates may comprise first to eighth two-input AND gates, an input of each of the first and fourth AND gates being connected to receive the first square wave signal, an input of each of the fifth and eighth AND gates being connected to receive the second square wave signal, an input of each of the second and third AND gates being connected to receive the third square wave signal and an input of each of the sixth and seventh AND gates being connected to receive the fourth square wave signal; the first sampling pulses being applied jointly to second inputs of the fifth and sixth AND gates, the second sampling pulses being applied jointly to second inputs of the third and fourth AND gates, the third sampling pulses being applied jointly to second inputs of the seventh and eighth AND gates and the fourth sampling pulses being applied jointly to second inputs of the first and second AND gates. First and second four-input OR gates are provided. The inputs for the first OR gate are coupled respectively to outputs of the first, third, fifth and seventh AND gates and the inputs of the second OR gate are coupled respectively to outputs of the second, fourth, sixth and eighth AND gates, outputs of the first and second OR gates being connected to the memory means. The present invention will now be described, by way of example, with reference to the accompanying drawings, wherein:- Figure 1 is a block schematic diagram of an embodiment of a radio receiver made in accordance with the present invention, Figures 2A and 2H show waveforms applicable to the operation of the receiver shown in Figure 1, Figure 3 is a circuit diagram of one embodiment of a hysteresis circuit. Figure 4 is a circuit diagram of a digital embodiment of a demodulator suitable for use in the receiver in accordance with the present invention, and Figures 5A to 5M show waveforms applicable to the operation of the receiver shown in Figure 4. In the drawings the same reference numerals have been used to identify corresponding features. Referring to Figure 1, the receiver comprises an antenna 10 for receiving FSK signals f c ±δ, where f c is a nominal carrier frequency and δ is the offset frequency for example 4.5 kHz for a data signal of 512 bits/second. These signals are supplied to first inputs of first and second mixers 12, 14. A local oscillator 16 generating a frequency f L , where ideally f c = f L , is connected to second inputs of the first and second mixers 12, 14. In the case of the mixer 14, a 90 degree phase shifter 18 is connected into the signal path between the local oscillator 16 and the mixer 14. The difference outputs of the first and second mixers are + δ and +δ phase shifted by -π/2 when the input signal is f c +δ, and -δ and -δ phase shifted by -π/2 when the input is f c -δ. In a non-illustrated arrangement of the receiver's front end, the phase shifter 18 is inserted into one of the signal paths between the antenna 10 and the first or second mixer 12 or 14, and the local oscillator 16 is connected directly to the second inputs of the mixers 12, 14. The quadrature related difference signals are filtered in low pass filters 20, 22 and are then hard limited in respective limiting amplifiers 24, 25 which provide the I and Q square wave signals. First and second signal multipliers 26, 27 are provided. The I and Q square wave signals are applied to the first inputs of the second and first multipliers 27, 26 respectively. These square wave I and Q signals are applied to respective pulse forming networks, for example C-R differentiators 28, 30, which provide pulses corresponding to each edge of the respective square wave signal. The time constant of each pulse forming network is short compared to the bit rate, for example one hundredth of the period of the square wave signal. The pulse sequences corresponding to the I and Q square wave signals are applied to the multipliers 26, 27, respectively. The outputs from the multipliers 26, 27 are applied to a subtracting stage 32 which effectively forms the sum of the signals. A bistable memory element 34, for example a hysteresis circuit, is connected to the output of the subtracting stage 32. The memory element 34 is followed by a glitch removing filter 36 and a fixed level bit slicer 38. The operation of the receiver shown in Figure 1 will now be described with reference to the waveforms shown in Figures 2A to 2H. Figures 2A and 2B respectively show the amplitude limited I and Q square wave signals, the amplitudes of which vary between ±1. The pulse sequences formed by the networks 28, 30 are shown in Figures 2C and 2D, respectively. Figures 2E and 2F show the outputs X,Y of the multipliers 26,27, respectively, and Figure 2G illustrates the output of the subtracting stage 32. The output of the memory element 34 is shown in Figure 2H and it will be noted that the polarity change from binary "0" to "1" occurs at the occurrence of the first positive pulse in the output from the subtracting stage 32. Since the memory element 34 changes state at the occurrence of the first pulse following any change of phase in the signal received at the antenna 10, it maintains a substantially constant dc output which is independent of the difference frequencies derived by the first and second mixers 12,14. Hence it is possible to determine the original data using the fixed level bit slicer 38. Further, the demodulator is inherently capable of demodulating signals at high data rates, up to the limit expected in an amplitude limited system, because it is responsive to a single edge change. As the system gives some inherent noise filtering, the filter 36 can, with advantage, be a digital filter, thus avoiding group delay problems which would occur with an analogue filter. Figure 3 illustrates a hysteresis circuit which can be used as the memory element 34. The hysteresis circuit comprises a flip-flop circuit comprising NPN transistors 40,42 having their base and collector electrodes cross-coupled as is conventional and their emitter electrodes connected to a current source 44. The load circuit for the transistor 40 comprises the series connection of a first resistor 46, the collector-emitter path of a NPN transistor 48 and a second resistor 50 between the V cc line and the collector of the transistor 40. A similar series connection of a third resistor 52, the collector-emitter path of a NPN transistor 54 and a fourth resistor 56 between the V cc line and the collector of the transistor 42 constitutes the load circuit for the transistor 42. The signal output from the subtracting stage 32 (Figure 1) is in reality a balanced output which in turn forms balanced inputs 58,60 applied to the base electrodes of the transistors 48,54, respectively. Balanced signal outputs 62,64 are derived from the collector circuits of the transistors 48,54,respectively In operation whenever the voltage difference applied to the transistors 48,54 exceeds a certain value, then the flip-flop changes state and remains in that state until an opposite voltage difference occurs. It will be noted from the illustrated hysteresis circuit that the circuit comprises resistors and transistors and there are no intentional reactive components which means that it is economical to implement compared to the cost of a filter used in British Patent Specification 2143386A, discussed in the preamble. Additionally a hysteresis circuit also provides other advantages over using a filter, which advantages have already been referred to. Figure 4 is a digital embodiment of the single edge demodulator. The demodulator assumes as its inputs amplitude limited square waves I, I , Q and Q . The circuit comprises four pairs of two input AND gates 66,67; 68,69; 70,71 and 72,73. The I signal is applied to AND gates 66,69, the I signal to gates 67,68, the Q signal to the gates 70,73 and the Q signal to the gates 71,72. Pulse sequences I', I ', Q' and Q ' are derived from the amplitude limited I and Q square wave signals using monostable circuits 74 to 77 and inverters 78,79 and are respectively applied to the AND gates 70,71; the gates 72,73; the gates 68,69 and the gates 66,67. Two four-input OR gates 80,81 are provided. Inputs A to D of the OR gate 80 are connected respectively to the outputs of AND gates 66,68,70 and 72 and inputs A to D of the OR gate 81 are connected respectively to the outputs of AND gates 67,69,71 and 73. Outputs E,F of the OR gates 80,81, respectively, are connected to a set/reset flip-flop formed by two two-input NOR gates 82,83, an input to one being the output of the other, and vice versa. Balanced outputs Z, Z are derived from the gates 82,83, respectively. The operation of the circuit shown in Figure 4 will now be described with reference to the waveforms shown in Figures 5A to 5M. Figures 5A and 5B show the I and Q amplitude limited square waves, respectively, the other square wave inputs I and Q being the antiphase versions of the I and Q signals, respectively. Figures 5C to 5F are the pulse sequences occurring at the outputs of the monostable circuits 74 to 77, respectively. It will be noted that the pulses in each sequence corresponding to the positive going edge in the respective square wave signal and that the duration of each pulse is small compared to that of the associated square wave. In the pairs of AND gates the input square wave signals are effectively sampled at a point in time phase shifted from their edges by substantially 90 degrees using the short duration pulses produced by the monostable circuits 74 to 77. Figures 5G to 5J show the inputs A to D of the OR gate 80 which inputs have been referenced A 80 , B 80 , C 80 , and D 80 . The inputs A to D of the OR gate 81 are not shown but can be deduced by examination of Figures 5A to 5F. Figures 5K and 5L illustrate the outputs E, F of the OR gates 80, 81, respectively, which outputs are applied to the set/reset flip-flop, the output Z of which is shown in Figure 5M. The operation of the embodiment shown in Figure 4 is generally equivalent to that of the analogue embodiment shown in Figure 1 in that the edge detection using the AND gates 66 to 73 is equivalent to the multipliers 26, 27 (Figure 1) and the subtracting stage 32 and bistable memory element 34 (Figure 1) are generally equivalent to the OR gates 80, 81 and set/reset flip-flop 82, 83.

Claims: A receiver for FSK signals, comprising means (10 to 22) for receiving the FSK signals and producing quadrature related frequency down-converted signals, means (24, 25) for amplitude limiting each of the frequency down-converted signals to form first and second square wave signals and demodulating means (26 to 34 or 66 to 83) for producing an output signal, characterised in that the demodulating means comprises means (26, 27, 28, 30 or 66 to 79) for sampling the first and second square wave signals at changes in polarity of the second and first square wave signals, respectively, and for producing respective sequences (X, Y) of pulses having a duration less than the period of said first or second square wave signal, means (32 or 80, 81) for combining the sequences (X, Y) of pulses to provide an output which changes polarity at the occurrence of the first pulse from the sampling means following a change of frequency in the received FSK signal and means (34 or 82, 83) for providing an analog output representative of the transmitted base band signal, said last means comprising memory means responsive to the occurrence of a change in polarity in the combined signal for producing a first substantially constant dc output until the next following change in polarity is detected, and responsive at the detection of said next following change in polarity for producing a second substantially constant dc output. A receiver as claimed in claim 1, characterised in that said means (26, 27, 28, 30 or 66 to 79) for sampling includes means (28, 30 or 74 to 79) for generating pulses having a duration of substantially one hundredth of the period of the first or second square wave signal. A receiver as claimed in claim 2, characterised in that said means for sampling comprises first means (26) for multiplying the second square wave signal by pulses generated in response to changes in polarity in both directions of the first square wave signal to produce a first sequence (X) of pulses and second means (27) for multiplying the first square wave signal by pulses generated in response to changes in polarity in both directions of the second square wave signal to produce a second sequence (Y) of pulses. A receiver as claimed in claim 1 or 2, characterised in that there is provided means for providing third and fourth square wave signals which are inverted versions of the first and second square wave signals, respectively, in that the means(74 to 79) for sampling comprises means for sampling the occurrence of alternate edges having the same polarity change in the first , second, third and fourth square wave signals and producing respective first to fourth sequences (I', I ',Q', Q ') of pulses and a plurality of logic gates(66 to 73) operative on the first to fourth square wave signals in response to said first to fourth sequences of pulses to provide further sequences of pulses which are applied to said combining means. A receiver as claimed in Claim 4, characterised in that the logic gates comprise first to eighth two-input AND gates (66 to 73), an input of each of the first and fourth AND gates (66, 69) being connected to receive the first square wave signal ( I ), an input of each of the fifth and eighth AND gates (70, 73) being connected to receive the second square wave signal (Q), an input of each of the second and third AND gates (67, 68) being connected to receive the third square wave signal (I) and an input of each of the sixth and seventh AND gates (71, 72) being connected to receive the fourth square wave signal ( Q ); the first sampling pulses ( I ') being applied jointly to second inputs of the fifth and sixth AND gates (70, 71), the second sampling pulses (Q') being applied jointly to second inputs of the third and fourth AND gates (68, 69), the third sampling pulses ( I ') being applied jointly to second inputs of the seventh and eighth AND gates (72, 73) and the fourth sampling pulses ( Q ') being applied jointly to second inputs of the first and second AND gates (66, 67) and in that first and second four-input OR gates (80, 81), are provided, inputs of the first OR gate (80) being coupled respectively to outputs of the first, third, fifth and seventh AND gates (66, 68, 70, 72) and inputs of the second OR gate (81) being coupled respectively to outputs of the second, fourth, sixth and eighth AND gates (67, 69, 71, 73), outputs of the first and second OR gates being connected to the memory means (82, 83). A receiver as claimed in any one of Claims 1 to 5, characterised in that the memory means (34) comprises a hysteresis circuit comprising nonreactive components. A receiver as claimed in any one of Claims 1 to 6, characterised in that a fixed level bit slicer (38) is coupled to the output of the memory means (34). A receiver as claimed in Claim 7, characterised in that a digital glitch removing filter (36) is coupled between the output of the memory means (34) and the fixed level bit slicer (38).

Inventors: LUFF GWILYM FRANCIS, WILSON JOHN FRANCIS, YOUELL RICHARD JOHN, LUFF, GWILYM FRANCIS, WILSON, JOHN FRANCIS, YOUELL, RICHARD JOHN, LUFF, GWILYM FRANCIS, C/O PHILIPS RADIO COM. SYST., WILSON, JOHN FRANCIS, C/O PHILIPS RADIO COM. SYST., YOUELL, RICHARD JOHN, C/O PHILIPS RADIO COM. SYST.
Applicants: PHILIPS ELECTRONICS NV, PHILIPS ELECTRONICS UK LTD, PHILIPS ELECTRONICS N.V., PHILIPS ELECTRONICS UK LIMITED
Application year: 1990
Main classification: H04L27
IPC codes: H04L27, H04B7, H03D3