A power supply fault protection circuit

Abstract: A MOSFET power switch (Q₁) in a DC-to-DC converter of a television apparatus has a drain electrode (D) that is coupled via a primary winding (T 1A ) of a chopper transformer (T₁) to an input supply voltage (B+). A first signal (V G ) at a given frequency causes, during normal operation, alternate conduction and nonconduction in the transistor (Q₁). A second signal (V Q2 ) is generated that is proportional to the voltage developed across the internal "on" resistance of the transistor (Q₁) when the transistor is conductive. The second signal (V Q2 ) is clamped when the MOSFET (Q₁) is nonconductive. The second signal (V Q2 ) is coupled via a current sensing arrangement (51) to the transistor gate (G). The second signal (V Q2 ) prevents the transistor (Q₁) from further conducting when the amplitude of the second signal (V Q2 ) exceeds a predetermined value that corresponds with a predetermined limiting threshold level of current (i₁) in the transistor (Q₁).

Description: The invention relates to a fault protection circuit in a power supply of, for example, a television apparatus and, in particular, to one that prevents an over-current condition in a power MOSFET transistor. EP-A-0 255 067 shows a circuit of this art. Within this circuit a measuring device in form of a differential amplifier is disabled when a switching power transistor is non-conductive. However, the signal to be measured still contains the large peaks which occur when the switching power transistor is non-conductive. As a result, the measuring is not as accurate as would be desired. A DC-to-DC converter, or chopper, used in, for example, a television apparatus power supply, may utilize a switching power transistor operating at a given frequency and having a first main current conducting electrode that is coupled to a first terminal of a primary winding of a transformer. A second terminal of the primary winding may be coupled to an input DC voltage that may be obtained from a battery or from a rectifier that rectifies an AC mains voltage. When the transistor is switched off, a voltage is induced in a secondary winding of the transformer. The induced voltage in the secondary winding is rectified for producing an output DC voltage that energizes a load circuit. The load circuit may be, for example, a deflection circuit of a television receiver. It may be desirable to protect the power transistor against an over-current condition that may be destructive. Such over-current condition may be caused by, for example, an excessive loading condition in the secondary winding of the transformer. In one prior art television circuit, a current level sampling resistor is coupled in series with the main current conducting electrodes of the power transistor. A voltage that is representative of the current flowing between the main current conducting electrodes is developed across the sampling resistor. A comparator, responsive to the voltage across the sampling resistor, generates an output signal when an over-current condition occurs. The over-current condition occurs when the current in the power transistor exceeds a predetermined threshold level that is considered a safe operating level. The comparator output signal is coupled to a drive circuitry of the power transistor to turn off the power transistor, thus preventing further conduction in the power transistor when the over-current condition is detected. In high current applications, such sampling resistor, disadvantageously, may dissipate considerable power. Hence, the efficiency of the converter is reduced. Therefore, it may be desirable to sense the current in the power transistor without interposing such sampling resistor in series with the power transistor. In a circuit embodying a feature of the invention, the power transistor is, for example, of the MOSFET (Metal Oxide Semiconductor Field Effect Transistor) type. The source electrode is grounded and the drain electrode is coupled to the primary winding of the aforementioned transformer. The voltage that is developed between the drain electrode and ground is proportional to the current in the MOSFET during its conduction. Therefore, such voltage is indicative of the level of the current that flows in the MOSFET during conduction. Such voltage is used for generating an over-current protection signal, embodying an aspect of the invention. On the other hand, the voltage between the drain electrode and ground, when the MOSFET is nonconductive, may be substantially higher and may not be indicative of the current that flows during the conduction time of the MOSFET transistor. It may be desirable to prevent that voltage from affecting the over-current protection signal. In accordance with an aspect of the invention, the voltage that is developed at, for example, the drain electrode, is used for generating a first portion of a first signal, embodying a feature of the invention. The voltage at the drain electrode may be referred to as the voltage that is developed across the "on" resistance of the MOSFET. When the MOSFET is conductive, the level of the first portion of the first signal varies proportionally as the voltage at the drain electrode. The level of the first portion of the first signal is indicative of the current in the transistor. When the MOSFET is nonconductive, a second portion of the first signal is established at, or clamped to a constant level such as, for example, zero volts. In this way, an amplitude of the first signal is, advantageously, indicative of the current in the transistor but is unaffected by the voltage at the drain electrode when the transistor is nonconductive. The first signal is coupled to a current sensing arrangement that may include a comparator. When a magnitude of the first signal exceeds a predetermined level, indicative of an over-current condition, the current sensing arrangement generates a control signal that is coupled to the gate electrode of the MOSFET, causing it to turn off. In this way, the MOSFET is advantageously protected against a potentially destructive over-current condition. In one embodiment of the invention, a second transistor is used for clamping the first signal to the zero volt level when the MOSFET power switch is nonconductive. A fault detection and protection apparatus, embodying an aspect of the invention, in a switching power supply includes a source of an input supply voltage and a first switching power transistor having first and second main current conduction electrodes and a control electrode. The apparatus includes a source of a first control signal at a given frequency that is coupled to the control electrode of the first switching power transistor. The first control signal causes, during normal operation, the first switching transistor to operate at a switching frequency determined in accordance with the given frequency. A utilization circuit of the apparatus includes an impedance coupled to the input supply voltage and to the first main current conducting electrode for developing an output supply voltage in the utilization circuit. The impedance forms a current path to a switched main current that flows in the first switching power transistor. A first current sensing signal is developed at a frequency determined in accordance with the given frequency between the first and second main current conducting electrodes of the first switching power transistor when the first switching power transistor is conductive. The first current sensing signal is indicative of a level of the main current when the first switching power transistor is conductive. A first portion of a second current sensing signal that is indicative of the level of the main current is generated at an output terminal, in accordance with the first control signal and in accordance with the first current sensing signal, when the first switching power transistor is at a conductive state. The output terminal is decoupled from the power transistor when the first switching power transistor is at a nonconductive state to prevent a voltage that is developed at the first main current conducting electrode of the first switching power transistor when the power transistor is nonconductive, from having a substantial contribution to an amplitude of the second current sensing signal. A second control signal is generated when a magnitude of the first portion of the second current sensing signal is outside a normal operation range, indicating an occurrence of a fault condition in the power supply. The second control signal is coupled to the control electrode of the first switching power transistor for preventing the first switching power transistor from further conducting. In the Drawing: FIGURE 1 illustrates a first embodiment of an over-current protection circuit, embodying an aspect of the invention; FIGURES 2a-2d illustrate waveforms useful in explaining the operation of the circuit of FIGURE 1; FIGURE 3 illustrates a second embodiment of an over-current protection circuit, embodying another aspect of the invention; FIGURES 4a-4d illustrate waveforms useful for explaining the operation of the circuit of FIGURE 3; and FIGURE 5 illustrates a third embodiment of an over-current protection circuit embodying a further aspect of the invention that is utilized in a push-pull output stage. FIGURE 1 illustrates a DC-to-DC converter, or chopper arrangement 100 of a television apparatus that includes an over-current protection arrangement 100a, embodying an aspect of the invention. FIGURES 2a-2d illustrate waveforms useful for explaining the operation of arrangement 100 of FIGURE 1. Similar numbers and symbols in FIGURES 1 and 2a-2d indicate similar items or functions. Arrangement 100 of FIGURE 1 includes an oscillator and driver stage 50 that generates a square wave signal V G at a given frequency during normal operation, as shown in FIGURE 2a. During normal operation, signal V G of FIGURE 1 causes a MOSFET Q₁ to switch on and off, alternately, at the switching frequency that is determined by that of signal V G . A primary winding T 1a of a chopper transformer T₁ has a first terminal that is coupled to a drain electrode of transistor Q₁ and a second terminal that is coupled to a DC input voltage B+. The switching operation of transistor Q₁ causes a voltage to be induced in a secondary windin T 1b of transformer T₁. Such voltage is rectified by a rectifier arrangement 60 comprising a diode D₁ and a capacitor C₁. Rectifier arrangement 60 produces a DC output voltage V RL across a load, such as a resistor R L . Resistor R L is representative of the load formed by circuitry of the television apparatus, not further illustrated in FIGURE 1, that is energized by voltage V RL . As a result of the conduction of transistor Q₁, a ramping current i₁ flows in winding T 1a , as shown in FIGURE 2b. The dashed lines in FIGURES 2b-2d correspond to a first example in which current i₁ is larger than in a second example, shown by a solid line. During, for example, interval t₁-t₂ of FIGURE 2c, current i₁ of FIGURE 1 develops an upramping voltage V Q1 , at the drain of transistor Q₁, that is proportional to current i₁. Voltage V Q1 is equal to the magnitude of current i₁ multipled by r DS , where r DS denotes the value of the "on" or forward resistance of transistor Q₁ when transistor Q₁ is fully turned-on. When transistor Q₁ is nonconductive, such as during interval t₂-t₁′ of FIGURE 2c, voltage V Q1 of FIGURE 1 is, as a result of transformer operation of transformer T₁, more positive than voltage B+. During interval t₁-t₂, for example, voltage V Q1 of FIGURE 2c is indicative of the magnitude of current i₁ of FIGURE 1. During interval t₂-t₁′ of FIGURE 2c, voltage V Q1 includes a portion that is not readily representative of current i₁ in transistor Q₁ of FIGURE 1. In accordance with an aspect of the invention, a current sensing signal V Q2 is generated from voltage V Q1 that is proportional to current i₁ when transistor Q₁ is conductive, and that is, for example, zero when transistor Q₁ is nonconductive. Therefore, signal V Q2 is, advantageously, unaffected by voltage V Q1 , during the "off" time of transistor Q₁. In this way, an amplitude of signal V Q2 , such as, for example, its peak-to-peak amplitude, is proportional to current i₁ that occurs during the "on" time of transistor Q1. Such feature of signal V Q2 is advantageous in that it simplifies further processing of signal V Q2 because signal V Q2 is unaffected by voltage V Q1 when transistor Q₁ is nonconductive. Such processing of signal V Q2 may include, for example, low-pass filtering signal V Q to reduce noise or averaging signal V Q2 to form a signal that is indicative of the average value or of the peak value of current i₁ of FIGURE 1. To generate signal V Q2 , the drain electrode of transistor Q₁ is coupled through a resistor R₁ to a drain electrode Q 2a of a JFET (Junction Field Effect Transistor) Q₂ operating as a shunt switch. The gate electrode of transistor Q₂ is coupled to signal V G . Transistors Q₁ and Q₂ operate in a complementary manner; when one is conductive, the other is not, and vice versa. Signal V Q2 , embodying an aspect of the invention and developed at electrode Q 2a , as shown in FIGURE 2d, is equal, during interval t₁-t₂, to voltage V Q1 of FIGURE 2c. During interval t₂-t₁′, for example, that occurs when transistor Q₁ is nonconductive, voltage V Q1 may not be readily representative of current i₁ of FIGURE 1. During interval t₂-t₁′, signal V Q2 of FIGURE 2d is clamped to approximately zero volts the "on" resistance of conductive JFET transistor Q₂ of FIGURE 1. Therefore, the contribution of voltage V Q1 to signal V Q2 when transistor Q₁ is conductive is substantially larger than when transistor Q₁ is nonconductive. Therefore, advantageously, a change in current i₁ will cause, proportionally, a larger change in, for example, the peak amplitude or the average of signal V Q2 than if signal V Q2 were not so clamped. The result is an over-current detection arrangement that is more sensitive to variations in current i₁. It should be understood that resistor R₁ and transistor Q₂ form a controllable attenuator having a very high or infinite attenuation factor that attenuates mainly when transistor Q₁ is nonconductive and that couples, for example, unattenuated voltage V Q1 to form signal V Q2 , when transistor Q₁ is conductive. Advantageously, signal V Q2 of FIGURE 2d is an analog signal. As an analog signal, signal V Q2 is capable of assuming any value from a range including infinite distinct values when current i₁ assumes any value from a corresponding range of values. In contrast, a digital signal may assume only a finite numbers of values. Being an analog signal, signal V Q2 of FIGURE 1 may be, advantageously, readily low-pass filtered in a conventional low-pass filter for obtaining a filtered current sensing signal, not shown in the FIGURES, that is coupled to current sensing arrangement 51. Note that if voltage V Q1 were low-pass filtered directly, its amplitude would not be readily representative of current i₁. This is so, because the portion of voltage V Q1 when transistor Q1 is nonconductive may not be necessarily related to current i₁ in magnitude. Such portion may be much larger than the other portion of voltage V Q1 when transistor Q₁ is conductive that makes the process of extracting information of current i₁ more difficult. Signal V Q2 is coupled to an input terminal 51a of current sensing arrangement 51 that may include the aforementioned low-pass filter. An over-current condition may occur when current i₁ of FIGURE 2b exceeds a predetermined level that is considered a safe operating level. When such over-current condition occurs, the peak or the average value signal V Q2 becomes larger than a corresponding predetermined reference voltage level. Consequently, current sensing arrangement 51 of FIGURE 1, that may include a comparator, generates an over current protection signal INHIBIT that is coupled to a control terminal 50a of oscillator and driver 50. Signal INHIBIT causes signal V G to be maintained at, for example, zero volts, that forces transistor Q₁ to remain nonconductive. Current i₁ of FIGURE 1 becomes zero, thereby protecting transistor Q₁ from the aforementioned over-current condition. The clamping operation of transistor Q₂, advantageously, simplifies the manner by which information regarding the magnitude of current i₁ is obtained from voltage V Q1 for producing signal INHIBIT. FIGURE 3 illustrates a power supply arrangement 100′, embodying another aspect of the invention, that is similar to arrangement 100 of FIGURE 1. In particular, the magnitude of the current in each MOSFET transistor, Q₁ and Q₁′ of FIGURES 1 and 3, respectively, is obtained from a voltage that is developed between the corresponding drain and source electrodes when the respective transistor is on. Similar numbers and symbols in FIGURES 1 and 3 indicate similar items or functions. A signal V S ′ of FIGURE 3 that is analogous to signal V Q2 of FIGURE 1 contains information of the magnitude of a current i₁′. FIGURES 4a-4d illustrate waveforms useful for explaining the operation of arrangement 100′ of FIGURE 3. Similar numbers and symbols in FIGURES 3 and 4a-4d indicate similar items or functions. In arrangement 100′ of FIGURE 3, a signal V G ′ that controls the switching operation of transistor Q₁′ is coupled to an anode electrode of a diode D₂′ via a resistor R₂′. Diode D₂′, operating as a series coupled switch, has a cathode electrode that is coupled to the drain electrode of transistor Q₁′. Signal V G ′ supplies a small forward biasing current through diode D₂′ at the same time that signal V G ′ causes transistor Q₁′ to turn on. Consequently, signal V s ′ is proportional to current i₁′. Signal V s ′ is developed at a terminal 51a′ that is coupled at a junction between resistor R₂′ and the anode of diode D₁′. Signal V s ′ is equal, during interval t a -t b of FIGURE 4d, to the magnitude of current i₁′ multiplied by resistance r DS ′ plus the level of voltage V D2 ′. Voltage V D2 ′ is the forward voltage drop across diode D₂′ of FIGURE 3 that is approximately equal to 0.7 volts. When signal V G ′ is zero, such as, for example, during interval t b -t c of FIGURE 4a, occurring when transistor Q₁ is nonconductive, the current in resistor R₂′ of FIGURE 3 is zero and diode D₂′ is back biased. The nonconductive diode D₂′ advantageously isolates input terminal 51a′ from voltage V Q1 ′. When diode D₂ is nonconductive, voltage V Q1 is higher than voltage B+. Thus, during, for example, interval t b -t c of FIGURE 4d, signal V s ′ is maintained at approximately zero volts. When signal V G ′ of FIGURE 3, due to some fault condition, is too small, or when the magnitude of current i₁′ is excessive, protection signal INHIBIT′ of current sensing arrangement 51′ is generated. Signal INHIBIT′, advantageously, prevents transistor Q₁′ from being turned on. In some switched mode power supply applications it may be desirable to increase the frequency of signal V G ′ when current i₁′ increases so as to increase the range of regulation of the DC output voltage in the face of corresponding variations in current i₁ and in voltage B+. Advantageously, signal V s ′ may also be coupled, as shown in FIGURE 3 by a dashed line, to a frequency controlling terminal 50b′ of oscillator and driver 50′ to increase, for example, the frequency of signal V G ′ when the magnitude current i₁′ increases, and vice versa. FIGURE 5 illustrates a converter 100″ embodying a further aspect of the invention. Converter 100″, that includes a push-pull stage, generates a 24 volt output voltage V₂₄ by DC-to-DC converting a 12 volt battery input voltage V B . Converter 100″ of FIGURE 5 may be used in, for example, a portable color television receiver having a ten inch picture tube of the type 250ATB22, made by Hitachi Co., and not shown in the figures. Additionally, converter 100″ of FIGURE 5 may provide a 15 volt standby supply voltage V SB and a 220 volt supply voltage V DG . Voltage V DG may be used for energizing a resonant degaussing circuit, not shown in the figures. In converter 100″, the drain electrodes of MOSFET'S Q₁₀ and Q₂₀ are coupled to end terminals 90 and 91, respectively, of a primary windin T 10a of a chopper transformer T₁₀. A terminal 92 of windin T 10a , located intermediate terminals 90 and 91, is coupled to battery voltage V B . An oscillator U₁ generates signals at opposite phases that are coupled via a drive circuit U₂ to form oppositely phased signals V 2a and V 2b , respectively. Signals V 2a and V 2b coupled, respectively, to the gate electrodes of transistors Q₁₀ and Q₂₀, via resistors R₃ and R₄, cause transistors Q₁₀ and Q₂₀ to operate in a push-pull manner. A drain current i₁₀ of transistor Q₁₀ is sampled by a diode D₂₀ in a manner that is analagous to that done by diode D₂′ of FIGURE 3. Consequently, a voltage V S10 of FIGURE 5 is formed in a similar manner to the way signal V S ′ of FIGURE 3 is formed. Similarly, a drain current i₂₀ of transistor Q₂₀ of FIGURE 5 is sampled by a diode D₃₀ to form a voltage V S20 . Voltages V S10 and V S20 , that are alternately developed, are applied to respective resistors R₆ and R₇. Consequently, a signal V SUM at a junction terminal A of resistors R₆ and R₇ is formed. Signal V SUM is developed across a resistor R18 and an integrating capacitor C6, and coupled to an inverting input terminal of a comparator U 3C . Diode D₂₀, for example, prevents voltage V₉₀ across transistor Q₁₀ from affecting signal V SUM when transistor Q₁₀ is nonconductive. Should a fault condition occur that causes either of currents i₁₀ and i₂₀ to exceed a predetermined level, signal V SUM will exceed a reference voltage V RF , developed across a diode D₆ at a noninverting input terminal of comparator U 3C . Consequently, a signal V₀ will be generated at an output terminal of comparator U 3C . Signal V₀ is coupled via a noninverting input terminal of a comparator U 3B having an output terminal that feeds an inverting input terminal of a comparator U 3A . As a result Signal V₀ occurs when signal V SUM is greater than reference voltage V RF . When signal V₀ is generated, a signal INHIBIT at a first or HIGH, logic level is generated at an output terminal of comparator U 3A . Signal INHIBIT at the first logic level causes the output signals of oscillator U₁ to remain at zero volts. The result is that each of signals V 2a and V 2b becomes zero. Therefore, the switching operation in transistors Q₁₀ and Q₂₀ ceases and transistors Q₁₀ and Q₂₀ are prevented from being turned on. Therefore, signal V SUM becomes zero and signal V₀ returns to its normal operation level. Comparator U 3B operates as a time-out timer. Once its output signal V 3BOUT changes, signal V 3BOUT is prevented from immediately returning to its normal operation level. Signal V 3BOUT is so prevented until a shut-down period of, illustratively, 150 milliseconds has elapsed from the time signal V₀ returns to its normal operation level. The 150 millisecond minimum delay is determined by the time constant that is associated with a capacitor C₅, coupled to an inverting input terminal of comparator V 3B , as described below. During normal operation of converter 100″, signal V SUM is smaller than voltage V RF . An output terminal of comparator U 3C , pin number 13, represents a high impedance that is formed at an open collector transistor, not shown in the figures. An output terminal of comparator U 3B , pin number 14, also represents a high impedance. Capacitor C₅ is charged to a voltage that is determined by resistors R₁₅, R₁₄, R₁₃ and R₁₆ that form a voltage divider. When signal INHIBIT at the output terminal of comparator U 3A is low, signal INHIBIT enables oscillator U₁. When an overload condition occurs, signal V SUM increases above the level of voltage V RF and triggers comparators U 3C and U 3B . Their output terminals switch to the LOW state that is approximately at ground potential and drive comparator U 3A output signal INHIBIT to the HIGH state level which disables oscillator U₁. As a result of converter 100″ being disabled, signal V SUM decreases to a value that is below that of voltage V RF and comparator U 3C output terminal switches to a high impedance. Because of the feedback provided by resistor R₁₃, comparator U 3B output signal V 3BOUT remains at the LOW state until capacitor C₅ discharges through resistor R₁₄ to a voltage that is lower than the voltage of signal V₀. Signal V₀ is developed at the noninverting input terminal of comparator U 3B . Consequently, comparator U₃B output terminal switches to the high impedance and capacitor C₅ begins charging via resistors R₁₆, R₁₃ and R₁₄. When the voltage at the inverting input terminal of comparator U 3A , pin number 6, has increased above the voltage at the noninverting input terminal, pin number 7, comparator U 3A output signal INHIBIT switches to the LOW state and again enables oscillator U₁. Therefore, the generation of signals V 2a and V 2b is resumed. The time interval between the triggering that is caused by signal V SUM and the time that the switching operation of transistors Q₁₀ and Q₂₀ ceases is, advantageously, very short, approximately 1 microsecond. After the switching operation in transistors Q₁₀ and Q₂₀ stops, the switching operation, cannot immediately resume, as indicated before. The resumption of the switching operation may occur, at the earliest, after the 150 millisecond shut-down period has elapsed, as described before. Thus, transistors Q₁₀ and Q₂₀ can recover or cool down from the over-current condition before they can be turned on again. In the circuit of FIGURE 5, the triggering level occurs when signal V SUM exceeds approximately 0.7 volts that is determined by the forward voltage of diode D₆. Such level corresponds to voltages V₉₀ and V₉₁ each being at approximately 0.8 volts. In each of MOSFET'S Q₁₀ and Q₂₀ of FIGURE 5, the "on" resistance that occurs when the transistor is fully turned on may be approximately 0.12 ohms. Thus, shut-down of converter 100″ occurs when the current i₁₀ or i₂₀ exceeds approximately 6.5 amperes. The protection threshold voltage or current may be controlled by changing the value of resistor R₁₈. Voltage comparator U₃ A receives, via its noninverting input terminal, a reference voltage developed in a zener diode D₁₆. When battery voltage V B falls below, for example, 11 volts, signal INHIBIT is generated because signal U 3BOUT at the inverting input terminal of comparator U 3A becomes smaller than the voltage at the noninverting input terminal V RF . Therefore, the switching operation of transistors Q₁₀ and Q₂₀ ceases. Consequently, further discharge of the battery is prevented. The positive portions of voltages V₉₀ and V₉₁ are coupled to a charge storage capacitor C₁₀ via diodes D₄ and D₅, respectively. Capacitor C₁₀ is thereby charged to approximately 24 volts. A voltage that is developed across a secondary winding T 10b of transformer T₁₀ is rectified and doubled in a voltage doubler 66 comprising a capacitor C₁₂, a diode D₈, a diode D₇ and a capacitor C₁₃. Voltage doubler 66 produces a voltage V DG that may be used as the supply voltage in a resonant degaussing circuit, not illustrated. Voltage V DG is also coupled to the gate of a MOSFET Q₃₀ to turn on the transistor. Capacitor C₁₀ is coupled via MOSFET Q₃₀ to a terminal 77 to develop voltage V₂₄ at terminal 77 when the switching operation is normal. When the switching operation ceases, as a result of, for example, an over-current condition, transistor Q₃₀ becomes immediately nonconductive. When transistor Q₃₀ becomes nonconductive, the transistor isolates terminal 77 from storage capacitor C₁₀. In this way, transistor Q₃₀, advantageously, immediately decouples power to the load.

Claims: Fault detection and protection apparatus in a switching power supply, comprising: a source of an input supply voltage (B+); a first switching power transistor (Q1) having first (D) and second (S) main current conducting electrodes and a control electrode (G); a source (50) of a first control signal (VG) at a given frequency that is coupled to said control electrode of said first switching power transistor (Q1) for causing, during normal operation, said first switching transistor (Q1) to operate at a switching frequency determined in accordance with said given frequency; a utilization circuit (T₁) including an impedance (T 1a ) coupled to said input supply voltage and to said first main current conducting electrode for developing an output supply voltage (VRL) in said utilization circuit, whereby said impedance (T1a) forming a current path to a switched main current (il) that flows in said first switching power transistor (Q1) to develop a first current sensing signal (VQ1) at a frequency determined in accordance with said given frequency, and that is developed between said first (D) and second (S) main current conducting electrodes of said first switching power transistor (Q1) when said first switching power transistor (Q1) is conductive, said first current sensing signal (VQ1) being indicative of a level of said main current when said first switching power transistor (Q1) is conductive; characterized by first means (Q2) responsive to said first control signal (VG) and to said first current sensing signal (VQ1) and having resistor (R1) that is coupled to said first main current conducting electrode (D) for generating at an output terminal (Q2a) a first portion of a second current sensing signal (VQ2) that is indicative of said level of said main current (i1) when said first switching power transistor (Q1) is at a conductive state, and for decoupling said output terminal (Q2a) from said power transistor (Q1) when said first switching power transistor (Q1) is at a non-conductive state to prevent a voltage (VQ1) that is developed at said first main current conducting electrode (D) of said first switching power transistor (Q1) when said power transistor is non-conductive from having a substantial contribution to an amplitude of said second current sensing signal (VQ2); and second means (51) responsive to said second current sensing signal (VQ2) for generating a second control signal (INHIBIT), when a magnitude of said first portion of said second current sensing signal (VQ2) is outside a normal operation range indicating an occurrence of a fault condition in said power supply, said second control signal (INHIBIT) being coupled to said source (50) of said first control signal (VG) controlling control said first switching power transistor (Q1) for preventing said first switching power transistor (Q1) from further conducting. Apparatus according to claim 1, characterized in that said first means comprises a second switching transistor (Q2) having a control electrode that is coupled to said first control signal (VG) and a main current conducting electrode that is coupled via said resistor (R1) to said first main current conducting electrode (D) of said first switching power transistor (Q1) for developing said second current sensing signal (VQ2) at said main current conducting electrode of said second switching transistor (Q2) such that when said first switching power transistor (Q1) is conductive said second switching transistor (Q2) is non-conductive, and vice versa. Apparatus according to claim 1, characterized in that said first means (Q2) operates at a frequency that is related to that of said first control signal (VG) and wherein, when said first switching power transistor (Q1) is non-conductive, said first means (Q2) causes said second current sensing signal (VQ2) to be at a predetermined constant level that is substantially lower than of said first portion of said second current sensing signal (VQ2). An apparatus according to Claim 1, characterized in that said first means comprises a diode (D2′) having a first terminal that is coupled to said first main current conducting electrode (D) of said first switching power transistor (Q1) for developing said second current sensing signal (VD2′) at a second terminal of said diode (D2′) such that, when said first switching power transistor (Q1′) is conductive, said diode (D2′) is forward biased, and when said first switching power transistor (Q1) is non-conductive, said first current sensing signal (VQ1′) that is developed at said first main current conducting electrode (D) of said first switching power transistor (Q1′) back biases said diode (D2′) to decouple said first main current conducting electrode (D) of said first switching power transistor (Q1′) from said second terminal (anode) of said diode (D2′). An apparatus according to Claim 1, characterized in that said impedance comprises, a first transformer winding (T10a) having a first terminal (90) that is coupled to said first main current conducting electrode (D) of said first switching power transistor (Q10), a second switching power transistor (Q20) having a first main current conducting electrode (D) that is coupled to a second terminal (91) of said first transformer winding (T10a) that is remote from said first switching power transistor (Q10) for conducting a second switching current in said second switching power transistor (Q20) and having a control electrode that is responsive to a control signal (V2b) at a frequency that is related to that of said first control signal (V2a) such that said first (Q10) and second (Q20) switching power transistors operate in a push-pull manner, and wherein said second means (51) is coupled to said first main current conducting electrodes (D) of said first (Q10) and second (Q20) switching power transistors for generating said second current sensing signal (VSUM) that is indicative of an overcurrent condition in either one of said first (Q10) and second (Q20) switching power transistors. An apparatus according to Claim 5, characterized in that said second means comprises first (D20) and second (D30) diodes, said first diode (D20) having a first terminal that is coupled to said first main current conducting electrode (D) of said first switching power transistor (Q10) and said second diode (D30) having a first terminal that is coupled to said first main current conducting electrode (D) of said second switching power transistor (Q20) and combining means (R6, R7) coupled to said diodes (D20,D30) for developing said second current sensing signal (VSUM) in accordance with a signal (VS10) that is developed at a second terminal of said first diode (D20) and in accordance with a signal (VS20) that is developed at a second terminal of said second diode (D30). An apparatus according to Claim 5, characterized by a rectifier arrangement (D4,D5) coupled to said first winding (T10a) for generating a first DC output voltage (VDG) in a capacitor (C10) and a switching third means (D30) responsive to a signal that is developed at a second winding (T10b) of said transformer (T10) for coupling said first DC output voltage to an output terminal (77) of said third means (Q30) during normal operation and for fast decoupling said output terminal (77) from said capacitor (C10) when a fault condition occurs. An apparatus according to Claim 1, characterized by a delay means (U3b) responsive to said second current sensing signal (VSUM) for generating a signal (V3BOUT) that enables resumption of the switching operation in said first switching power transistor (Q10) only after a period having at least a predetermined length has elapsed following the occurrence of said second control signal (INHIBIT) that is sufficient to prevent harm to said first switching power transistor (Q10).

Inventors: ASCHWANDEN FELIX, BART THEODOR ERNST, HAFERL PETER EDUARD, ASCHWANDEN, FELIX, BART, THEODOR ERNST, HAFERL, PETER EDUARD
Applicants: RCA THOMSON LICENSING CORP, RCA THOMSON LICENSING CORPORATION
Application year: 1989
Main classification: H02H7
IPC codes: H04N5, H02H7, H02M3