Voltage controlled oscillator free from free-run oscillating frequency adjustment

Abstract: A voltage controlled oscillator free from free-run oscillating frequency adjustment is disclosed. This oscillator includes an amplifier having a positive feed-­ back loop, a resonator connected to the output of the amplifier, and a variable reactance circuit connected to the output of the amplifier. The reactance circuit presents a reactance value variable by a control voltage to control an oscillation frequency of the oscillator. There is a further provided a low-pass filter which is connected in cascade to the variable reactance circuit. A high-pass filter is preferably inserted in the positive feedback loop.

Description: BACKGROUND OF THE INVENTION Field of the Invention The present invention relates to a voltage controlled oscillator (called hereinafter a "VCO") and, more particularly, to a VCO free from adjustment of a free-run oscillating frequency. Description of the Prior Art A VCO has been widely used for various systems. The VCO has a free-run oscillating frequency which should be adjusted to match operation of the system in which the VCO is used, such as PLL (Phase Locked Loop). For such usage, it is preferable that the free-run oscillating frequency is automatically set to a center frequency of a capture range of the PLL circuit without any adjustment. In the prior art, the VCO used in the PLL circuit employs a resonance circuit consisting of a resistor and a capacitor, the value of the resistor and/or the value of the capacitor being adjusted to bring the free-run oscillating frequency into conformity with the center frequency. Such adjustment requires considerable man-hours and time. Therefore, a VCO free from the free-run oscillating frequency adjustment has been widely demanded. One of such adjustment free VCO in the prior art is shown in Fig. 1 in a block diagram. This VCO includes an amplifier 8, a ceramic resonator 10 and a variable reactance circuit 11. The output of the amplifier 8 is positively fed back to the input thereof, and further connected to the ceramic resonator 10 and the variable reactance circuit 11. The output of the amplifier 8 is led out from an output terminal 9 of the VCO. The equivalent capacitance of the variable reactance circuit 11 is changed in accordance with a control voltage applied thereto via a control terminal 12. This equivalent capacitance will be referred to as "C v ". As shown in Fig. 1, the ceramic resonator 10 can be expressed by an equivalent circuit consisting of a parallel capacitance Ca, a series capacitance Cr, a series inductance L and a series resistance r. Therefore, the oscillation frequency f o obtained from the output terminal 9 can be expressed by the following formula (1): The oscillation frequency f o of the VCO is thus controlled by changing the equivalent capacitance Cv of the variable reactance circuit 11 (i.e., by the control voltage applied to the control terminal 12). Moreover, the variable reactance circuit 11 is designed in such a manner that its capacitance Cv becomes zero at the time of free-run oscillation of VCO. In other words, at the time when no control voltage is applied to the control terminal 12, the free-run oscillating frequency of VCO is automatically determined at the resonance frequency inherent to the ceramic resonator 10. Thus, the adjustment of the free-run oscillating frequency is unnecessary by employing the resonator 10 whose resonance frequency is the above-mentioned center frequency. Here, it is noted that the capacitance Cv of the variable reactance circuit 11 is designed at zero when no control voltage is applied but is designed so as to change from a negative value to a positive value by the control voltage. Thus, the VCO shown in Fig. 1 is free from the adjustment of the free-run oscillating frequency. However, this VCO frequently oscillates at an abnormal oscillation frequency locked at a transient time such as a power switching-on. More specifically, the ceramic resonator 10 has structurally two resonant frequencies, as seen from the impedance-to-frequency characteristic shown in Fig. 2. That is, the VCO oscillates stably at both of a first frequency f₁ of a lower resonant frequency and a second frequency f₂ in an upper resonant frequency. The second frequence f₂ is as large as about 10 times than the first frequency f₁. As is understood from the formula (1), when the equivalent capacitance component Cv of the variable reactance circuit 11 approaches to the parallel capacitance Ca, the oscillation frequency f o becomes higher to approach to the second frequency f₂. In the normal operation, the variable range of the equivalent capacitance Cv of the variable reactance circuit 11 is controlled not to approach to the parallel capacitance Ca. However, at the transient response time such as power switching-on, this control does not operate sufficiently, so that the VCO is locked to oscillate at the oscillation of the second frequency f₂. Moreover, the oscillation loop including the ceramic resonator 10 does not often operate, so that the oscillation inherent to the circuit structure of the amplifier 8 and the variable reactance circuit 11 occurs. This oscillation frequency is lower than the frequency f₁. In any of these cases, the VCO does not oscillate at the desired free-run oscillation frequency, so that the PLL circuit does not operate. Patent Abstracts of Japan, Vol. 8, No. 29 and US-A-4.587.500 discloses VCO's with tuning of a resonance circuit (cooperating with a positive-feedback-loop amplifier) via a variable reactance circuit. SUMMARY OF THE INVENTION An object of the present invention is to provide an improved VCO free from free-run oscillating frequency adjustment. Another object of the present invention is to provide a VCO free from adjustment of a free-run oscillating frequency, while preventing abnormal free-run oscillation. A VCO according to the present invention comprises a voltage controlled oscillator oscillating at a frequency which is variable by a control voltage, said oscillator comprising an amplifier having input and output terminals and a positive feedback circuit provided between the input and output terminals, a resonator coupled to the output terminal of said amplifier, and characterised by circuit means supplied with said control voltage and having a controllable capacitance component which is controlled by said control voltage, and a low-pass filter coupled between said circuit means and the output terminal of said amplifier. That is the VCO according to the present invention is featured in that the low-pass filter is provided in the variable capacitance circuit. The low-pass filter prevents abnormal oscillation at the second resonant frequency of the resonator. A high-pass filter may be provided in the positive feedback loop of the amplifier to prevent the VCO from oscillating at a frequency determined only by the amplifier and the capacitance circuit. BRIEF DESCRIPTION OF THE DRAWING The above and other objects, advantages and features of the present invention will be more apparent from the following description taken in conjunction with the accompanying drawings, in which: Fig. 1 is a block diagram representative of a prior art VCO; Fig. 2 is a diagram representative of an impedance-to-frequency characteristic of a ceramic resonator; Fig. 3 is a block diagram representative of a VCO according to an embodiment of the present invention; and Fig. 4 is a circuit diagram of a PLL stereo demodulator employing a VCO according to another embodiment of the present invention. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Referring to Fig. 3, a VCO 50 according to an embodiment of the present invention includes a differential amplifier 1, a high-pass filter 4, a ceramic resonator 5, a variable reactance circuit 6 having a los-pass filter 7 therein, and a bias voltage source 8. The output node of the amplifier 1 is led out as an oscillation output terminal 2 and is fed back to a non-inverting input node (+) thereof through the high-pass filter 4. That is, the amplifier 1 has a positive feedback loop provided with the high-pass filter 4. The inverting input node (-) of the amplifier 1 is biased by the bias voltage source 8. The ceramic resonator 5 and the variable reactance circuit 6 including the low-pass filter 7 are connected to the output terminal 2. In other words, the circuit 6 and the low-pass filter 7 are connected in cascade to the amplifier 1. An oscillation control voltage is supplied to the circuit 6 via a control terminal 3. As shown in the equivalent circuit diagram of Fig. 3, the ceramic resonator 5 consists equivalently of a parallel capacitance Ca, a series capacitance Cr, a series inductance L and a series resistance r connected as shown. The cut-off frequency of the low-pass filter 7 is designed to be between the first and second resonant frequencies f₁ and f₂ of the ceramic resonator 5 shown in Fig. 3, and the cut-off frequency of the high-pass filter 4 is designed to be lower than the first resonant frequency f₁. Accordingly, the oscillation frequency f o of the VCO 50 of this embodiment is also expressed by the formula (1). Upon the free-run oscillation, i.e. upon non-controlling, the variable reactance circuit 6 is constructed such that the equivalent capacitance Cv thereof is zero. Therefore, the free-run oscillating frequency of the VCO 50 is equal to the resonance frequency inherent to the ceramic resonator 5, so that the adjustment of the free-run frequency is not required. The control voltage applied to the terminal 3 changes the capacitance Cv of the variable reactance circuit 6 from a negative capacitance value to a positive capacitance value. The oscillation frequency f o of the VCO 50 is thereby controlled. Since the cut-off frequency of the low-pass filter 7 is designed to be between the first and second resonant frequencies f₁ and f₂ of the resonator 5 to cut off the frequency higher than the frequency variable range required to the VCO 50. Namely, the low-pass filter 7 restricts the equivalent capacitance Cv of the variable reactance circuit 6 to change in a negative direction. As a result, the oscillation at the second resonant frequency f₂ of the ceramic resonator 5 is prevented. The high-pass filter 4 disposed in the positive feedback loop of the amplifier 1 cuts off the frequency lower than the frequency variable range required to the VCO 50. Therefore, it is possible to prevent oscillation at the inherent frequency which is determined solely by a capacitance component in the amplifier 1 and the variable reactance circuit 6. Thus, to VCO 50 is controlled to make the most stable oscillation at the frequency near the first resonant frequency f₁ of the ceramic resonator 5 by the low-pass filter 7 and the high-pass filter 4, and can be free from the abnormal oscillation locked at the transient time such as a power switching-on. Turning to Fig. 4, there is shown a PLL stereo demodulator circuit employing a VCO 50′ according to another embodiment of the present invention. An FM composite signal containing a sum signal (R+L) of right (R) and left (L) channels, a difference signal (R-L) modulated by a 38 KHz subcarrier signal and a 19 KHz pilot signal is supplied to one of two input terminals of a phase comparator 40, the other of which is supplied with a 19 KHz signal from a second frequency divider 42 described later. The phase comparator 40 compares the phase of the signal from the frequency divider 42 with that of the pilot signal in the FM composite signal and generates the D.C. voltage signals representative of the phase difference therebetween at the output terminals 41 and 42 in a true and complementary manner. That is, when no FM composite signal is supplied to the phase comparator 40, i.e. when the VCO 50' is free-running, the terminals 41 and 42 have the same voltage level as each other. When the FM composite signal is supplied, the terminal 41 changes from that voltage level in a positive or negative direction in accordance with the phase difference, whereas the terminal 42 changes in an opposite direction by the same amount. The output voltage signal from the phase comparator 40 controls the oscillation frequency of the VCO 50'. The oscillation signal of VCO 50' is frequency-divided by a first frequency divider 51 to produce a 38 KHz signal. This 38 KHz signal is divided to obtain the 19 KHz signal by the second frequency divider 52. The 38 KHz signal from the frequency divider 51 is supplied to a stereo demodulator 43 which is also supplied with the FM composite signal. VCO 50′ includes an amplifier 1′, a bias voltage source 8′, a high-pass filter 4′, a ceramic resonator 5′, and a variable reactance circuit 6′ having a low-pass filter 7′, similarly to the VCO 50 shown in Fig. 3. The amplifier 1′ includes transistors Q₁₁ and Q₁₂ connected in a differential form, a current source I₁₁, and three pairs of transistors Q₁₃ and Q₁₄; Q₁₅ and Q₁₆; and Q₁₇ - Q₁₈, each pair forming a current mirror. The base of the transistor Q₁₂ operates as an inverting input terminal (-) of the amplifier 1′ and is thus supplied with a bias voltage from the bias source 8′ through a resistor R₁₂. The connection point of the transistors Q₁₄ and Q₁₈ is an output terminal 2′ of the amplifier 1′ and is thus fed back to the base of the transistor Q₁₁ (the non-inverting input terminal (+) of the amplifier 1′) through the high-pass filter 4′. The bias source 8′ is also connected to the non-inversion input terminal (+) through a resistor R₁₁. The high-pass filter 4′ consists of a capacitor C₄₁ and a resistor R₄₁. The ceramic resonator 5′ and the variable reactance circuit 6′ are connected to the output terminal 2′. The variable reactance circuit 6 includes transistors Q₆₁ through Q₆₇, current sources I₆₁ through I₆₅, a capacitor C₆₁ and a resistor R₆₁ which are connected as shown. Particularly, the output 2′ of the amplifier 1′ is supplied to the base of the transistor Q₆₂ through the emitter-follower transistor Q₆₇ and the low-pass filter 7′ consisting of a resistor R₇₁ and a capacitor C₇₁. The transistor Q₆₁ is connected to the transistor Q₆₂ through the capacitor C₆₁ in a differential form. The base of the transistor Q₆₁ is connected to the bias source 8′ through the resistor R₆₁. The first output terminal 41 of the phase comparator 40 is connected in common to the bases of the transistors Q₆₃ and Q₆₆, and the second output terminal 42 is connected in common to the bases of the transistors Q₆₄ and Q₆₅. The collectors of the transistors Q₆₃ and Q₆₅ are connected in common to the collector of the transistor Q₁₅ and the collectors of the transistors Q₆₄ and Q₆₆ are connected in common to the collector of the transistor Q₁₃. The transistors Q₆₃ and Q₆₅ constitutes a double balance differential circuit. In the construction described above, the phase comparator 40 generates the same voltage at the output terminals 41 and 42 upon the free-running, and hence the currents flowing through the transistors Q₆₃ to Q₆₆ are all equal to one another. Accordingly, the equivalent capacitance of the variable reactance circuit 6′ is zero, so that the VCO 50′ oscillates in a free-running mode at the first resonant frequency f₁ of the ceramic resonator 5′. In this embodiment, the free-run oscillating frequency is designed to be 456 KHz. The second resonant frequency f₂ of the ceramic resonator 5' is about 4 to 6 MHz. The resonator 5' has the capacitance Ca of about 300 pF. The capacitance value of the capacitor C₇₁ and the resistance value of the resistor R₇₁ in the low-pass filter 7' are 16 pF and 10 KΩ, respectively. The cut-off frequency of the low-pass filter 7' is thus about 1 MHz. Therefore, it is possible to prevent the oscillation of the VCO 50' at the second resonant frequency f₂ (4 to 6 MHz). Since the free-run oscillating frequency of VCO 50' is 456 KHz, the frequency division ratio of the frequency divider 51 is set to 1/12, thereby the 38 KHz signal is derived therefrom. The phase comparator 40 compares the phase of the input pilot signal and the signal from the frequency divider 52 and controls the base voltage difference between the transistors Q₆₂ (Q₆₆) and Q₆₄ (Q₆₅) in accordance with phase difference. Since the output is taken out of the variable reactance circuit 6' in the double-end fashion, the equivalent capacitance Cv can be made variable from -2C₆₁₁ to +2C₆₁₁, where C₆₁₁ represents the capacitance value of the capacitor C₆₁. In this embodiment, the capacitance value C₆₁₁ is 100 pF. As a result, the oscillation frequency of VCO 50' is controlled so that there occurs a predetermined phase difference between the pilot signal and the signal from the frequency divider 52. The stereo demodulator 43 subjects the FM composite signal to the stereo demodulation by use of the 38 KHz signal from the frequency divider 51 and thus generates the right channel signal R and the left channel Signal L, as well-known in the art. The values of the capacitor C₄₁ and resistor R₄₁ in the high-pass filter 4' are set to 33 pF and 15 KΩ, respectively. Therefore, the high-pass filter 4' has the cut-off frequency of about 100 Hz, and thus prevent the VCO 50' from oscillating at the inherent frequency determined only by the amplifier 1' and the variable reactance circuit 6' (about 50 Hz). The present invention is not limited to the above embodiment, but may be modified and changed without departing from the scope of the invention as defined by the claims. For example, the ceramic resonator 5 (5') may be replaced by a crystal resonator.

Claims: A voltage controlled oscillator oscillating at a frequency which is variable by a control voltage, said oscillator comprising an amplifier (1) having input and output terminals and a positive feedback circuit (4) provided between the input and output terminals, a resonator (5) coupled to the output terminal of said amplifier, and characterised by circuit means (6) supplied with said control voltage and having a controllable capacitance component which is controlled by said control voltage, and a low-pass filter (7) coupled between said circuit means and the output terminal of said amplifier. The oscillator as claimed in Claim 1, wherein said resonator has a first resonant frequency and a second resonant frequency different from said first resonant frequency and said low-pass filter (7) has a cut-off frequency between said first and second resonsant frequencies. The oscillator as claimed in Claim 1, wherein said circuit means includes first and second transistors (Q11,Q12) connected in a differential form through a capacitor and said low-pass filter (R₇₁,C₇₁) is connected between the output terminal of said amplifier and a control electrode of said first transistor. The oscillator as claimed in Claim 1 comprising a high-pass filter (4) in the positive feedback circuit from the output terminal to the input terminal of said amplifier. The oscillator as claimed in Claim 4, wherein said resonator has a first resonant frequency and a second resonant frequency higher than said first resonant frequency, said high-pass filter having a cut-off frequency lower than said first resonant frequency, and said low-pass filter having a cut-off frequency between said first and second resonant frequencies. The oscillator as claimed in Claim 4, wherein said amplifier has an inverting input terminal (-) and a non-inverting terminal and said high-pass filter (4) is coupled between said output terminal and said non-inverting input terminal, said inverting input terminal being supplied with a bias voltage.

Inventors: OHKUBO FUMIKAZU NEC IC MICROCO, OHKUBO, FUMIKAZU NEC IC MICROCOMPUTER SYSTEM, LTD.
Applicants: NIPPON ELECTRIC CO, NEC CORPORATION
Application year: 1989
Main classification: H03B5
IPC codes: H03B5