Squaring circuits in mos integrated circuit technology

Abstract: A squaring circuit (FIG. 1, 100; FIG. 3, 300) includes an operational amplifier (10) having its positive input terminal connected to ground, its negative input terminal connected through a nonlinear voltage-to-current conversion device (FIG. 1, T₂, T₃; FIG. 3, M₂, M₃, T₂) to a balanced source of the input voltage (±V IN ), and its output terminal connected through a linear current-to-voltage conversion device (FIG. 1, R₁: FIG. 3, M₁, C₁) to its negative input terminal. The circuit can be made to have a balanced output (FIGS. 2 and 5), and the circuit can be made in the configuration of either a continuous-time circuit (FIGS. 1 and 2) or a sampled data circuit (FIGS. 3 and 5).

Description: In telecommunication systems, squaring circuits are useful in a variety of contexts. For example, since squaring a wave changes its frequency in a signal receiver, a squaring circuit is useful for shifting the frequency range ("band") of an incoming modulated carrier wave to a different frequency range. Such shifting is useful in many situations. In particular, it is useful in modems at a receiver where it is desired to process the information on a pulsed carrier wave emanating from a transmission channel in which the information is coded as symbols represented by means of phase modulation of the pulses on the carrier. That is, the carrier is modulated by the (typically non-rectangular) pulses, and these pulses are periodic except for a separate phase shift that is imposed upon each pulse in accordance with the symbol corresponding to that pulse, whereby the pulse contains information. In such cases, among other things, it is desired at the receiver to determine the symbol rate-i.e., to achieve "timing recovery"--by first squaring the incoming modulated carrier wave by means of a squaring circuit in order to produce or enhance the energy component at the symbol rate prior to further signal processing, the phase modulated pulsed carrier itself as it emanates from the channel containing little or no energy at the symbol rate. As is known in the art, it is important in these cases that the output of the squaring circuit be pure. That is, it is important that the output produced by the squaring circuit be free of all odd-power terms, i.e., be free of all frequency components that have frequencies equal to odd-integer multiples of the input frequency, as well as be free of all even-power terms of higher order than the second. Otherwise, confusion would be introduced by the squaring circuit into the determination ("recovery") of the symbol rate. In U.S. Patent No. 4,585,961, issued on April 29, 1986, to S. J. Dauben entitled "Semiconductor Integrated Circuits for Squaring a Signal with Suppression of the Linear Component," a squaring circuit having a pure output is disclosed. The circuit contains a pair of matched substantially identical paths to which an input voltage is applied Each of the paths consists of a double-ended difference amplifier, which receives the input voltage, and a dual-to-single-ended converter, which receives the output of the difference amplifier and delivers its own single-ended output to a nonlinear/summing device--such as a squarer/summer network--for nonlinearly transforming and summing the single-ended outputs of the coverters in the two paths. Although the Daubert squaring circuit is useful for many purposes, nevertheless, because its MOS transistors operate in saturation and hence its output contains a relatively large d.c. offset (a relatively large term of zero order), an added voltage level shifter is needed in those cases where the output of the squaring circuit is used to drive another circuit in which this offset is undesirable, for example, another circuit in which all signals are referenced to ground, as in switched-capacitor filters and continuous-time filters. It would be desirable to have a squaring circuit in MOS technology without a dc offset in the output and hence without the need for the added level shifter, because the added shifter consumes an undesirable amount of semiconductor chip area. According to this invention there is provided a squaring circuit as claimed in claim 1. In some specific embodiments (FIGS. 1 and 2) the circuit has a configuration that is useful for continuous-time operation; in other specific embodiments (FIGS. 3 and 5) the circuit has a sampled-data configuration. In a specific embodiment (FIG.3) of the sampled data configuration, the need for matched (substantially identical) transistors is avoided. Also, in some specific embodiments (FIGS. 2 and 5) the circuit has a balanced output configuration, which is useful for improved performance in that a balanced circuit in general has the advantages of improved power supply noise rejection and larger signal handling capability. The invention will now be described with reference to the accompanying drawings, in which: FIG. 1 is a schematic diagram of a squaring circuit embodying the invention and having a continuous-time operational configuration and a single-ended output configuration; FIG. 2 is a schematic diagram of a squaring circuit embodying the invention and having a continuous-time operational configuration and a balanced output configuration; FIG. 3 is a schematic diagram of a squaring circuit embodying the invention and having a sampled-data configuration and a single-ended output configuration; FIG. 4 is a timing diagram useful in describing the operation of the circuit shown in FIG. 3; and FIG. 5 is a schematic diagram of a squaring circuit embodying the invention and having a sampled-data configuration and a balanced output configuration. As shown in FIG. 1, a continuous-time squaring circuit 100, having a circuit output terminal 14, includes a matched pair of N-MOS (n-channel MOS) transistors T₂ and T₃. By the term"matched pair" it is meant that the structures of the two transistors are substantially identical. The transistors T₂ and T₃ have one of their high-current carrying terminals (source or drain) connected to a common node N₁, and have another of their highcurrent carrying terminals connected to receive from a balanced source (not shown) a pair of balanced input voltages V IN and -V IN , respectively. As known in the art, where a system supplies only V IN , then -V IN can be obtained from a single-ended-to-double-ended voltage converter. Thus, a balanced source of V IN and -V IN is readily available. The transistors T₂ and T₃ (FIG. 1) have their gate electrodes biased to a common voltage V G and their substrates biased to another common voltage V B . For example, V G = 1.5 volts, and V B = -1.5 volts. A single-ended operational amplifier 10 (FIG. 1) has a negative input terminal 11 connected to the node N₁ and a positive input terminal 12 connected to ground (0 volts). An output terminal 13 of the amplifier 10 is connected through a resistor R₁ to the circuit output terminal 14 and to the node N₁. Advantageously, this resistor R₁ is linear, i.e., is characterized by a linear relationship between current through it and voltage across it; and thus the resistor R₁ serves to convert a current into a voltage in a linear manner. The circuit output terminal 14 is connected to an input terminal 161 of a utilization circuit 16, such as a filter circuit, for utilizing an output V OUT of the squaring circuit 100. It is important that the bias voltage V G and the input voltage V IN are selected such that the MOS transistors T₂ and T₃ operate in their non-saturation regions; that is, as known in the art, advantageously the peak-to-peak input V IN is maintained at values less than 2(V G -V T ) where V T is the threshold voltage, typically about 1.0 volt. The drain-to-source current I of each transistor operated in its non-saturation region is related to the source and drain voltages v S and v D , respectively, by the known expression: I = K [a₁(v D -v S ) + a₂(v 2 D -v 2 S ) + a₃(v 3 D -v 3 S ) + a₄(v 4 D -v 4 S ) + ..], where K is a constant that depends upon the size (channel width and length) of the transistor while a₁, a₂, a₃ and a₄ are constants that depend upon other parameters such as gate voltage V G , substrate voltage V B , and the remaining transistor structure parameters. It is important that T₂ and T₃ form a matched pair, so that they have the same K as well as the same a i for all i. It should be noted that in Eq. (1) it is assumed that v D and v S are measured with respect to ground, and that Eq. (1) can be found in a paper by M. Banu and Y. Tsividis entitled "Detailed Analysis of Nonidealities in MOS Fully Integrated Active RC Filters Based on Balanced Networks", published in IEE Proceedings , Vol. 131, Pt. G, No. 5, pp. 190-196 (Oct. 1984), at p. 190, Eq. (3). Advantageously, as is obtained with the ordinary MOS transistor, the properties of transistors T₂ and T₃ are such that the second-order term, a₂ (v 2 S -v 2 D ), is much larger than any of the other terms of higher power. Note that unlike the situation in the saturation region, in the non-saturation region there is no undesired zero-order (constant, or DC) term in the current I. During operation of the circuit 100, the transistors T₂ and T₃ operate in their non-saturation regions and deliver currents I₂ and I₃, respectively, to the node N₁. Since an operational amplifier cannot pass current through itself, the sum total of I₂ + I₃ = I₁ passes through the resistor R₁ which converts this current I₁ into a voltage drop I₁R₁. However, the operational amplifier 10, as is the case with any operational amplifier, for all practical purposes cannot support any voltage difference across its input terminals 11 and 12. But its positive input terminal 12 is grounded. Accordingly, the voltage at node N₁ will be essentially ground (0 volts), and the voltage drop I₁R₁ will be equal in absolute value to the output voltage V OUT which is developed at the output terminal 14. Thus, (I₂ + I₃)R₁ = I₁R₁ = -V₁ = -V OUT . On the other hand, assuming V IN to be positive, with T₂ and T₃ being substantially identified n-channel MOS (N-MOS) transistors, the drain voltage V D of T₂ will be equal to V IN , and its source voltage V S will be zero, again because the voltage at node N₁ will be equal to zero (ground). At the same time, the source voltage V S of transistor T₃ will be equal to -V IN , and its drain voltage V D will be zero. The drain-to-source currents of T₂ and T₃, each in accordance with Eq. (1), will be I₂ and -I₃, respectively. Hence, I₂ and -I₃ have equal but opposite even-order terms (±Ka₂V 2 / IN; ±Ka₄V 4 / IN;...) in their dependence (eq. 1) upon source and drain voltage, but have equal odd-order terms, provided only that the transistors T₂ and T₃ are matched so that they have the same K, as well as the same a i for all i. Consequently, all odd-order terms cancel one another in the expression I₁ = I₂ + I₃, and thus they do not appear in the output, whereas the even-order terms are doubled in the output: V OUT = -2R₁K(a₂V 2 IN + a₄V 4 IN + ...) Ordinarily the fourth-order term a₄V 4 IN , as well as all other higher order terms, is so small relative to the second-order term a₂V 2 / IN that it can be neglected for all practical purposes. Thus, according to Eq. (3), the squaring circuit 100 accurately produces an output voltage that is equal to the square of the input voltage to within a multiplicative constant (-2R₁Ka₂), i.e., is linearly proportional to the square of the input voltage, as is desired in a squaring circuit. The resistor R₁ in practice is not ordinarily easy to obtain as a linear device as assumed above. Moreover, the constant (-2R₁Ka₂) undesirably varies with semiconductor processing variations, because the resistor R₁ does not track the transistors T₂ and T₃. Further, the circuit 100 is not balanced in its output even though it is balanced in its input. Despite these considerations, the circuit 100 is useful in some contexts. FIG. 2 shows the squaring circuit 200, a balanced form of the circuit 100 described above with the resistor R, replaced by a transistor T₁. Elements of FIG. 2 that are similar to elements of FIG. 1 are denoted by the same reference labels. A utilization circuit 26, typically a balanced circuit, is connected for receiving balanced outputs V OUT and -V OUT which are developed at the squaring circuit output terminals 14 and 15. By inspection, the squaring circuit 200 in FIG. 2 replaces the single-ended operational amplifier 10 of the circuit 100 of FIG. 1 by a balanced output operational amplifier 20 and adds N-MOS transistors T₄, T₅, and T₆ on the positive input side of the amplifier, corresponding to N-MOS transistors T₁, T₂, and T₃, respectively, on the negative input side. Note that the input voltage applied to both T₅ and T₆, however, is ground potential rather than V IN and -V IN applied to T₂ and T₃. The positive and negative output terminals of the amplifier 20 are separately connected to circuit output terminals 14 and 15, respectively. Advantageously, the transistors T₅ and T₆ are matched to each other and to the transistors T₂ and T₃ (which are also substantially identical to each other). Likewise, the transistors T₁ and T₄ are mutually matched, but are not ordinarily matched to T₂, T₃, T₅, and T₆. The voltages V OUT and -V OUT which are developed at the circuit output terminals 14 and 15 are delivered to a pair of input terminals 261 and 262 of a utilization means 26. The voltages V OUT and -V OUT will be essentially the square and the negative of the square, respectively, of the input voltage V IN , as can be seen from the following analysis of operation. During operation, as explained above, the positive and negative input terminals of the operational amplifier 20 will have essentially the same voltage potential, say V x . Due to matching of transistors, K₂ = K₃ = K₅ = K₆ and K₁ = K₄. Accordingly, from Eq. (1): I₂ = K₂[a₁(V IN -V x ) + a₂(V 2 IN -V 2 x ) + a₃(V 3 IN -V 3 x )+ ...] -I₃ = K₂[a₁(V x +V IN ) + a₂(V 2 x -V 2 IN ) + a₃(V 3 x +V 3 IN ) + ...] where it should be noted again that in Eq. (5) the source-to-drain current flows through T₂ from right to left, whereas FIG. 2 indicates that I₂ flows from left to right in FIG. 2 (again assuming T₂ is an N-MOS transistor matched to T₃ and V IN is of positive polarity). Thus, since the current I₁, through T₁ is given by I₁ = I₂ +I₃, it follows that -I₁ = 2K₂[a₁V x + a₂(V 2 x -V 2 IN ) + a₃(V 3 x ) + ...] But it is also true from Eq. (1) applied to the transistor T₁ that: +I₁ = K₁[a₁(V x -V OUT ) + a₂(V 2 x -V 2 OUT ) + a₃(V 3 x -V 3 OUT ) + ...] Combining Eqs. (6) and (7): 2K₂[a₁V x + a₂(V 2 x -V 2 IN ) + a₃V 3 x + ...]= -K₁[a₁(V x -V OUT ) + a₂(V 2 x -V 2 OUT ) + a₃(V 3 x -V 3 OUT ) + ...] Similarly: -I₅ = K₂[a₁V x + a₂V 2 x + a₃V 3 x + ...] -I₆ = K₂[a₁V x + a₂V 2 x + a₃V 3 x + ...] +I₄ = K₁[a₁(V x + V OUT ) + a₂(V 2 x -V 2 OUT ) + a₃(V 3 x + V 3 OUT ) + ...] Likewise, since I₄ = I₅ + I₆, it follows that -K₁[a₁(V x + V OUT ) + a₂(V 2 x -V 2 OUT ) + a₃(V 3 x + V 3 OUT ) + ...]= 2K₂[(a₁V x + a₂V 2 x + a₃V 3 x + ...)] Combining Eqs. (8) and (12), it follows that -2K₂a₂V 2 IN -K₁[a₁(V x + V OUT ) + a₂(V 2 x -V 2 OUT ) + a₃(V 3 x + V 3 OUT ) + ...]= -K₁[a₁(V x -V OUT ) + a₂(V 2 x -V 2 OUT ) + a₃(V 3 x -V 3 OUT ) + ...], from which it follows that -K₂a₂V 2 IN = K₁[a₁V OUT + a₃V 3 OUT + ...]. Retaining only terms of up to second order -- i.e., assuming all terms of higher order than the second are negligible -- it follows from Eq. (14) that at positive output terminal of the amplifier 20, the output voltage will be given by V OUT = -(K₂a₂/K₁a₁)V 2 IN . as is desired in a squaring circuit. The balanced output amplifier 20 then ensures that the voltage -V OUT developed at the negative output terminal of the amplifier is indeed the negative of the foregoing expression for V OUT . Each of K₂ and K₁ in that ratio K₂/K₁, all other things being equal, depends on transistor device channel length and width dimensions, so that the required matching K₂ = K₃ = K₅ = K₆ and K₁ = K₄ can be achieved very accurately -- i.e., better than to within one percent -- and to track with variation in processing parameters so that K₂/K₁ remains the same. The circuits 100 (FIG. 1) and 200 (FIG. 2) are thus continuous-time squaring circuits in that they operate on continuous-time inputs and provide continuous-time outputs. Both these circuits require closely matched transistors, T₂ and T₃ in particular, any mismatch of these transistors can result in an undesired presence of odd-power terms, due to their non-exact cancellation, in the output V OUT . Moreover, in the circuit 200 the transistors T₁ and T₄ should be matched. These requirements of matching, however, can be avoided by using the sampled-data squaring circuit 300 as depicted in FIG. 3. FIG. 3 shows a sampled-data squaring circuit 300. Those elements in FIG. 3 which are similar to those in FIG. 1 are denoted by the same reference labels. Here in FIG. 3 the transistors M₁, M₂, M₃, and M₄ are clocked by clock pulse sequence ₁, ₂, ₃, and ₄, respectively. Each of M₁, M₂, M₃, and M₄ can be made in C-MOS, that is each of these transistors is an n-channel MOS transistor with a clock pulse sequence applied to its gate electrode, connected in parallel with a p-channel MOS transistor, with a complementary corresponding clock pulse sequence applied to its gate electrode. That is, each of the transistors M₁, M₂, M₃, and M₄ form a clocked C-MOS transmission gate where M₁ is clocked by the clock pulse sequence ₁ (and  ₁), M₂ by ₂ (and  ₂), M₃ by ₃ (and  ₃), and M₄ by ₄ (and  ₄). As known in the CMOS art, the n-type tub for the p-channel transistors is biased at V DD , typically +5 volts; the p-type tub for the n-channel transistors is biased at ground. The timing of the clock pulse sequences is shown in FIG. 4. Advantageously, the clock sequences ₁, ₂, ₃, ₄ have mutually non-overlapping pulses, and hence these sequences are mutually non-overlapping as to the time intervals during which they turn on MOS transistors. The clocked transistor M₄ acts as a sampler of the voltage at the output terminal 13 of the operational amplifier 10, whereas the clocked transistors M₂ and M₃ act as samplers of input voltages -V IN and V IN , respectively. Note that the resistor R₁ (FIG. 1) is replaced in the circuit 300 by a capacitor C₁ (FIG. 3) connected in parallel with the clocked transistor M₁. The capacitor C₁, in conjunction with the clocked transistor M₁, acts as an integrator of the current I₁ except during time intervals when ₁ turns on the transistor M₁ -- that is, during intervals t₀t₁, t₈t₉, etc. In this way, the capacitor C₁ converts the current I₁ to a voltage V₁ across the output and input terminals 13 and 11 of the operational amplifiers: as more fully discussed below. The input voltage V IN itself is assumed in FIG. 4 to have jumps at times t₀, t₈, etc., -- i.e., at the beginning of each clock cycle defined by the rising edge of ₁ -- but this is not essential; but it is always preferred than V IN should not change during the time interval t₂t₅, i.e., during the time interval starting with ₂ going high and ending with ₃ going low, that is, M₂ turning on and M₃ off. As noted below, it is also important that the time intervals t₂t₃ and t₄t₅ be of equal duration -- i.e., t₂t₃ = t₄t₅. During operation when ₁ goes high and thus turns on M₁ (while M₂, M₃, and M₄ are off) -- i.e., at time t₀ and then throughout the time interval t₀t₁ -- the capacitor C₁ is discharged to 0 volts across it. Thereafter, when ₂ goes high and thus turns on M₂ -- i.e., at t₂ and throughout t₂t₃ -- the current I₁ flows from -V IN through M₂ and T₂ into C₁. Advantageously, M₂ (as well as M₃) has a much lower on resistance than the (steady) resistance of T₂, so that the voltage drop across M₂ is negligible. Thus, the drain-to-source current -I₁ throughout the interval t₂t₃ is found by inspection of Eq. (1) to be equal to: -I₁ = K₂(+a₁V IN -a₂V 2 IN + a₃V 3 IN -a₄V 4 IN -...). Meanwhile during t₂t₃ the capacitor C₁ integrates this current I₁ to produce voltage V₁ across the output and input terminals 13 and 11 of the operational amplifier 10 as shown in FIG. 4. Again, the voltage at input terminal 11 will be essentially zero (ground) because the positive input terminal 12 of this amplifier is grounded. Thereafter, when ₃ goes high and thus turns on M₃ -- i.e., at t₄ and throughout t₄t₅ -- the current I₁, flows from +V IN through M₃ and T₂ into C₁, so that the current I₁ throughout the interval t₄t₅ is given by I₁ = +K₂(a₁V IN + a₂V 2 IN + a₃V 3 IN + a₄V 4 IN +....). Meanwhile during t₄t₅ the capacitor integrates this current I₁. Note that, in this integration during t₄t₅, the odd-power terms in I₁, including the linear term a₁V IN , thus will cancel the contributions of odd-power terms in I₁ obtained earlier from the integration during t₂t₃, provided that V IN essentially does not vary but remains essentially the same during t₂t₃ as during t₄t₅, and that these time intervals have equal duration: t₂t₃ = t₄t₅. Thus, at t₅ the voltage V₁ will contain only the even-power terms, i.e., essentially only the square term as desired, and hence V₁ is then valid at the output. Accordingly, subsequently during t₆t₇ the clock sequence ₄ goes high and thus turns on M₄ to enable the voltage V₁ at the output terminal 13 of the amplifier 10 to appear at the circuit output terminal 14 as V OUT . This output terminal 14 is connected to an input terminal 361 of a utilization circuit 36 for utilizing V OUT . Notice that, because the transistor T₂ time-shares the currents from V IN and -V IN , proper operation of the circuit 300 for squaring the input does not require any transistor matching. All that is required is a low on-resistance for both M₁ and M₂. The sampled-data squaring circuit 300 (FIG. 3) is not balanced but develops its single-ended output voltage V OUT at its sole circuit output terminal 14 for delivery to a utilization circuit 36. The sampled-data squaring circuit 300 can be modified, however, so as to become a balanced sample data squaring circuit 500, as depicted in FIG. 5. Elements in FIG. 5 that are similar to these depicted in FIG. 3 are denoted by the same reference labels. Here in FIG. 5, a mirror image (except for ground input) of the circuitry of FIG. 3 is added. Elements in the mirror image that correspond to the elements of FIG. 3 are denoted by the same subscript reference numerals plus ten. It is important that the transistor T₂ and T₁₂ are matched, and also that the capacitors C₁ and C₁₁ are also matched. A balanced output (V OUT , -V OUT ) is thus delivered to input terminals 561 and 562 of a utilization circuit 56. The circuit 500 (FIG. 5) operates relative to the circuit 300 (FIG. 3) in much the same way as the circuit 200 (FIG. 2) operates relative to the circuit 100 (FIG. 1), except that the capacitors C₁ and C₁₁ in the circuit 500 are linear devices for converting (charge and hence) current into voltage whereas T₁ and T₄ (FIG. 2) are nonlinear. Typically, the gate bias V G in the circuits 300 and 500 is equal to 3.5 volts, rather than 1.5 volts as in the circuits 100 and 200, while the substrate bias V B is -1.5 volts in the circuits 300 and 500, just as in the circuits 100 and 200. By varying the gate bias V G in any of the circuits 100, 200, 300, and 500 the constant K and hence also the magnitude of V OUT can be modified. Although the invention has been described in detail in terms of specific embodiments, various modifications may be made. For example, the transistors T₁ and T₄ in FIG. 2 may be replaced by matched resistors. The transistors M₁, M₂, M₃, and M₄ (FIG. 3) can be made in N-MOS or P-MOS instead of C-MOS technology; and the transistors T₁, T₂, T₃, T₄, T₅, T₆ can be made in P-MOS instead of N-MOS technology.

Claims: A squaring circuit (100; 200; 300; 500) comprising (a) an operational amplifier (10;20) having a positive (12) and a negative (11) input terminal and at least a first output terminal (13); (b) first linear current-to-voltage conversion means (R₁;T₁;M₁C₁) connecting the first output terminal (e.g., 13) of the operational amplifier to a negative input terminal (11) thereof, for supplying negative feedback to the amplifier, and (c) first nonlinear voltage-to-current conversion means (T₂T₃;M₂M₃T₂) connected to receive first and second (V IN ,-V IN ) balanced input voltages to be squared by the circuit, for delivering a current (I₁) to the first linear current-to-voltage conversion means, whereby the voltage (V OUT ) developed at the first output terminal (13) of the operational amplifier is essentially linearly proportional to the square of the input voltage with substantially no offset. A circuit as claimed in claim 1 wherein the nonlinear voltage-to-current conversion means comprises an MOS transistor (T₂) plus first and second clocked transistors (M₂M₃) which are clocked to be on during mutually non-overlapping periodic first and second successive time intervals (t₂t₃, t₄t₅), respectively, of mutually equal time duration during each clock cycle, the first and second clocked transistors (M₂M₃) being connected to receive the first and second input voltages (V IN ,-V IN ), respectively, for passing the first input voltage (V IN ) during the first time interval (t₂t₃) of each cycle and the second input voltage (-V IN ) during the second time interval (t₄t₅) of each cycle to the MOS transistor (T₂) operating in its non-saturation region. A circuit as claimed in claim 2 wherein the first linear current-to-voltage conversion means comprises a capacitor (C₁) connected in parallel with a third clocked transistor (M₁) which is clocked to be on during a third time interval (t o t₁) preceding the first time interval (t₂t₃) during each clock cycle, the third time interval (t o t₁) being nonoverlapping with respect to both the first (t₂t₃) and the second (t₄t₅) time intervals. A circuit as claimed in claim 3 comprising a utilization circuit (16;26;36;56) connected for utilizing the output voltage developed at the first output terminal of the amplifier. A circuit as claimed in claim 1 comprising second linear current-to-voltage conversion means (T₄;M₁₁C₁₁) connecting a second output terminal of the amplifier and a positive input terminal thereof, for supplying a second feedback to the positive input terminal; and second nonlinear voltage-to-current conversion means (T₅T₆;M₁₂M₁₃T₁₂) connected to a reference voltage terminal (ground), for nonlinearly converting the reference voltage into a second current (I₄), and connected to deliver the second current to the second current-to-voltage conversion means. A circuit as claimed in claim 5 wherein the first and the second nonlinear voltage-to-current conversion means comprises a first and a second pair, respectively, of first and second clocked transistors, each of the first clocked transistors (M₂M₃) being clocked to be on during a first time interval (t₂t₃) of each clock cycle, each of the second clocked transistors (M₁₂,M₁₃) being clocked to be on during a second time interval (t₄t₅) following the first time interval (t₂t₃) of each such cycle, the first and second time intervals being mutually non-overlapping and having mutually equal time durations, the first and the second clocked transistors of the first pair being connected to receive the first and second input voltages, respectively, for periodically passing these input voltages to a first MOS transistor (T₂) operating in its non-saturation region, and the first and second clocked transistors of the second pair both being connected to receive the reference voltage for passing the reference voltage to a second MOS transistor (T₁₂) operating in its non-saturation region. A circuit as claimed in claim 6 wherein each of the first and second linear current-to-voltage conversion means comprises a separate capacitor (C₁, C₁₁) connected in parallel with a separate third clocked transistor (M₁, M₁₁) which is clocked to be on during a third time interval (t o t₁) during each clock cycle, the third time interval (t o t₁) preceding the first time interval (t₂t₃) and being nonoverlapping with both the first and the second time intervals (t₂t₃,t₄t₅).

Inventors: KHOURY JOHN M, TROSINO JAMES M, KHOURY, JOHN M., TROSINO, JAMES M.
Applicants: AT & T CORP, AT&T CORP.
Application year: 1989
Main classification: G06G7
IPC codes: G06G7, H04L27