Accessing memory cells

Abstract: A memory cell (1) is read by first charging a pair of bit lines (13,14) to given positive potentials and then raising the potential of a cell access line (12) to render access transistors (10,11) conductive. The cell supply voltage (+V) is sufficient to cause substantial hot-electron stress in the n-channel transistors (3,5,10,11) of the cell if it were applied directly across their source-drain paths while they were conductive. However a limit is imposed on the maximum positive potentials which are applied to the bit lines (13,14) from the exterior, and on the minimum ratio of the sizes of the cell n-channel amplifier transistors (3,5) to the sizes of the access transistors (10,11) taking into account the threshold voltages of the amplifier transistors, and as a result substantial hot-electron stress does not occur. Substantial hot-electron stress is also prevented during a write operation by arranging that this is effectively preceded by a read operation.

Description: This invention relates to a memory comprising a CMOS submicron Static Random Access Memory (SRAM) cell, said cell comprising a pair of cross-coupled inverters, each inverter comprising a series connection of an inverter-PMOS-transistor, and an inverter-NMOS-transistor between two supply terminals, a supply voltage being applied across said two supply terminals in operation, outputs of the inverters being coupled to respective bit lines via respective NMOS-access-transistors, the memory further comprising precharging means for precharging said bit lines to a predetermined precharging voltage before executing a read operation. A memory as defined above is disclosed in an article in IEEE Journal of Solid-State Circuits, Vol. SC-22, No. 5, October 1987 entitled "A 40-ns/ 100 pF Low-Power Full-CMOS 256 K (32k x 8) SRAM", by W.C.H. Gubbels et al. This article discusses an advanced SRAM which is constructed by means of a 1.3µm process. Reliable transistors for a 5 V power supply are obtained using a lightly doped drain (LLD) structure for protecting the transistors against "hot electron stress". The hot-electron stress phenomenon is well-known per se, and is the result of charge carriers in the channel of the relevant transistor being accelerated by the source-to-drain electric field to sufficient energy to ionise the channel material on impact, so that valance band electrons of the channel material are excited into the conduction band. Each time such an event occurs the result is an extra conduction electron and a hole, which may flow to the drain and substrate respectively, giving rise to increased drain-source current and substrate current. If charge carriers generated by impact ionisation have sufficient energy, they are able to surmount the energy barrier between the channel material and the insulating material for the transistor gate. Once in the insulating material some carriers become trapped, creating both fixed charge in the insulator and interface traps, which in turn give rise to a change in the transistor characteristics. A present trend is a desire for memories having larger and larger storage capacities, and this implies a desire for a reduction in memory cell size in order that the larger number of cells required can be accommodated on a reasonably-sized area of substrate. A reduction in memory cell size entails a reduction in size of the constituent transistors of the cell. However, it has been found that if the channel lengths of the cell transistors are reduced to below 1µm (submicron), for example to 0.8 µm while the operating conditions are maintained unchanged "hot-electron stress" occurs in the NMOS transistors in spite of using a lightly doped drain (LLD) structure for the transistors, resulting in a drastically reduced life expectancy for the cell. It will be evident that the hot-electron stress could be avoided if the voltages supplied to the cells were suitably reduced but it has been found that a general such reduction results in a considerable increase in the time required to access the cells. It is an object of the invention to mitigate this disadvantage. The invention provides a memory as defined in the first paragraph which is characterized in that said supply voltage is sufficient, when applied directly across the channel of each individual inverter-NMOS-transistor and NMOS-access-transistor while that transistor is conductive, to result in hot-electron stress in that transistor, said precharging voltage is sufficiently below said supply voltage so as to prevent hot electron stress from occurring in the access-transistors, and the inverter-NMOS-transistor of each of the inverters and its associated NMOS-access-transistor having dimensions such as to keep a voltage representing a low logical level, when such a voltage present on a respective output of one inverter is transmitted to the corresponding bit line, below a safety value that is sufficiently below a threshold voltage of the inverter-NMOS-transistor of the other inverter to prevent hot electron stress from occurring in said inverter-NMOS-transistor of the other inverter. (It should be noted that the term "CMOS" is used herein in the sense in which it is commonly employed at the present day, and in consequence is to be understood to embrace complementary transistor structures in which the transistor gate electrodes are composed of non-metallic materials, for example polysilicon). The present invention is based upon the insight that in order to avoid the "hot electron stress" the maximum permissible voltage across the conduction channel of a NMOS-transistor depends on the voltage difference between the gate and source of the NMOS-transistor. Said maximum permissible voltage is larger for a voltage difference between the gate and source of the NMOS-transistor that lies below the safety value than a voltage difference between the gate and source of the NMOS-transistor that lies above the safety value. In an embodiment said supply voltage is substantially 5 Volts, said precharging voltage lies between substantially 2.5 Volts and 4 Volts, and said safety value is equal to substantially said threshold voltage minus 0.3 Volts. It has been found that the above-mentioned voltage range for the precharging level and the specified safety value can prevent the occurrence of hot electron stress in NMOS-transistors which are constructed by means of a submicron process (e.g. 0.8 µm). In a further embodiment the memory comprises control means for executing each write operation in such manner that it is preceded by precharging. Thus, in order to prevent hot electron stress from occurring in the access-transistors, the control means precharge the bit lines (e.g. to a voltage level typically higher than 2.5 Volts (e.g. 4 V) in the case of a 5 Volts power supply) prior to a write operation. In effect the voltage occurring across the access-transistors is then substantially less than the supply voltage. Thereupon the control means turn on the access-transistors, after this the control means put the desired information on the bit lines to write the memory cell and thus the respective outputs of the inverters in the memory cell take over the information on the respective bit lines. In a further embodiment the memory comprises control means for executing each write operation in such manner that it is preceded by a read operation. The desired precharging of the bit lines and selecting of the access-transistors prior to the write operation as discussed in the preceding paragraph can be accomplished by a read operation. An embodiment of the invention will now be described, by way of example, with reference to the accompanying diagrammatic drawings, in which:- Figure 1 is the circuit diagram of a static random access memory cell, Figure 2 is a diagram showing what has been found to be a "hot-electron safe" operating area for a particular construction of n-channel insulated-gate field-effect transistor structure, and Figure 3 is a circuit diagram of a static random access memory arrangement which includes cells as shown in Figure 1. In Figure 1, a static random access memory cell 1 comprises first and second CMOS inverting amplifier structures 2, 3 and 4, 5 respectively which are cross-coupled so as to form an arrangement which has a first stable state in which the potential at the output 6 of the first structure is positive relative to the potential at the output 7 of the second structure, and a second stable state in which the reverse is the case. It will be noted that, as shown, the amplifier structures are double cross-coupled, diodes 8 and 9, shown in dashed lines, being present between the p-channel transistor 2 and the n-channel transistor 3, and between the p-channel transistor 4 and the n-channel transistor 5 respectively. These diodes may or may not be present in practice. The gate electrodes of further n-channel insulated-gate field-effect transistor structures 10 and 11 are connected to a cell access signal supply conductor 12, and their channels connect the outputs 6 and 7 to first and second information signal conductors 13 and 14 respectively. The CMOS amplifier structures 2, 3 and 4, 5 are each connected between power supply conductors 15 and 16, conductor 15 carrying a positive potential V relative to conductor 16 which it will be assumed is at ground potential. As is known, when it is required that the information content of the cell 1 be read out onto the information signal conductors or bit lines 13 and 14, a positive potential can be applied to the cell access signal supply conductor 12 to render the further or access transistors 10 and 11 conductive, the sign of the resulting potential difference between the lines 13 and 14 then being indicative of which bistable state is currently present. Normally the lines 13 and 14 are precharged to a positive potential, for example +V, before the access transistors 10 and 11 are rendered conductive, in order to prevent an accidental write operation in the cell when this conduction occurs. When a write operation is actually required suitable different potentials, for example +V and zero respectively, the relative sign of which corresponds to the particular bistable state required can be impressed on the lines 13 and 14 and a positive potential can be applied to the line 12 to render the access transistors conductive, thereby setting the cell into the appropriate state if is it not already in this state. If the exemplary voltages quoted above are employed, and V=+5 volts ±10%, then the cell described operates satisfactorily if the channel lengths of the n-channel and p-channel transistors of the cell are 1.2 µm and 1.4 µm respectively as in the cell described in the article quoted in the preamble. However, if these channel lengths, particularly these of the n-channel transistors, are substantially reduced, for example to 0.8 µm, it is found that, when the exemplary voltages quoted above are employed and V=5 volts ±10%, substantial hot-electron stress occurs in the n-channel transistors, with a concomitant reduction in their life expectancy. A reduction in V to a value such as to prevent the substantial hot-electron stress results in a very considerable increase in the cell access time. In an exemplary embodiment of the invention, however, in which the channel lengths of the n-channel transistors are reduced to 0.8 µm, the value of V is maintained at +5 volts, but the maximum positive voltage which is applied to the conductors 13 and 14 from the exterior is limited to +4 volts ±10% and the threshold voltages of the transistors 3 and 5, together with the ratios between the sizes (channel conductances) of the transistors 3 and 5 and those of the transistors 10 and 11 respectively are chosen in a particular way which will be discussed below with reference to Figure 2. It has been found that, when this is done, substantial hot-electron stress in the n-channel transistors can be avoided without entailing the very considerable increase in cell access time which would occur if the potential difference between the conductors 15 and 16 were itself reduced. The diagram of Figure 2 shows what has been found to be the "hot-electron safe" working area of a particular n-channel insulated-gate field-effect transistor, this being the area below the full line 17. Drain-source voltage Vds has been plotted as the ordinate against gate-source voltage Vgs along the abscissa. It has been assumed that the maximum drain-source voltage possible is 5.5 volts, i.e. volts ±10%. It will be seen that the transistor is "hot-electron safe" when its gate-source voltage is above its threshold voltage T provided that its drain-source voltage is limited to 4.0 volts under these conditions (region 20). On the other hand, the transistor is "hot-electron safe" when its drain-voltage is 5.5 volts only if its gate-source voltage does not exceed a value which is 0.3 volts below the threshold voltage T (region 19). There is an intermediate transition region 18 between these two situations. (In fact there is not a sharp dividing line between what is, and what is not, "hot-electron safe". For example in some circumstances it may be permissible to increase the transistor gate-source voltage to within 0.25 volts of the threshold voltage T with a drain-source voltage of 4 volts before the resulting hot-electron stress reaches an impermissible magnitude. Moreover, the 4 volt level of the right-hand portion of the line 17 may in some circumstances be subject to a tolerance of 10%). When, therefore, the n-channel transistors in the cell of Figure 1 conform to the diagram of Figure 2, and the potential difference V between the conductors 15 and 16 is 5.5 volts, the operating points of these transistors must be kept below the line 17, or at least the line 17 after it has been subjected to the tolerance and condition relaxations quoted above. That this is in fact the case, provided that in accordance with the invention the maximum positive voltages on the conductors 13 and 14 are suitably limited and the threshold voltages of the transistors 3 and 5 together with the sizes of these transistors relative to the sizes of the transistors 10 and 11 are suitably chosen, will now be demonstrated. Consider first the quiescent (not selected) state of the cell of Figure 1, and assume for the sake of example that the cell is in a stable state in which output 6 of the amplifier 2, 3 is positive relative to the output 7 of amplifier 4, 5. In the quiescent state the line 12 is at ground potential so that transistors 10 and 11 are cut-off, isolating the cross-coupled inverting amplifiers 2, 3 and 4, 5 from the conductors 13 and 14. Hence within the cell the logic levels are CMOS logic levels, i.e. Vcc and ground. On the assumption that the threshold voltages of the n-channel transistors 3, 5, 10 and 11 are all about 1 volt, transistors 10, 11 and 3 operate in the low Vgs region 19 of Figure 2 whereas transistor 5 operates in the low Vds region 20. Consider now a read operation for the cell of Figure 1. According to an aspect of the invention the conductors are pre-charged to a positive potential not exceeding 4 volts while the potential on line 12 is maintained at a value such that transistors 10 and 11 are cut off. After this the potential on line 12 is raised in a positive direction, rendering transistors 10 and 11 conductive. Assuming that lines 13 and 14 are precharged to actually +4 volts then, as the potential on line 12 is raised towards +V current will start to flow from line 14 through transistors 11 and 5. This will raise the voltage at output 7. The operating regions of the n-channel transistors are as follows. Transistors 5 and 10 operate in the low Vds region 20 of Figure 2, as does transistor 11 because the voltage on conductor 14 is limited to 4 volts. Transistor 3 operates in the low Vgs region 19 of Figure 2 provided that the voltage at output 7 does not exceed (T-0.3) volts (or (T-0.25) volts if such relaxation is permissible). According to the invention the ratio between the channel conductances of the transistors 11 and 5, and the threshold voltage T of transistor 3, are chosen so that this requirement is satisfied, a corresponding choice also being made for the ratio between the channel conductances of the transistors 10 and 3, and the threshold voltage of the transistor 5. Typically the channel widths of transistors 5 and 3 may be chosen to be at least twice the channel widths of the transistors 10 and 11 respectively, if the maximum voltage on the conductors 13 and 14 is 4 volts and the threshold voltages T of the transistors 3 and 5 are 1.0 volt. Consider now a write operation for the cell of Figure 1. According to an aspect of the invention, in effect a read operation as described above is first carried out, after which the lines 13 and 14 are made relatively low and relatively high respectively (assuming that a change in the state of the cell is required) while the potential on line 12 is such that transistors 10 and 11 are conductive. Although theoretically it is possible that the drain-source voltage of transistor 10 will exceed 4 volts under these circumstances, in practice this will not be the case since the RC time constant of output 6 will in practice be sufficiently smaller than the RC time constant of the line 13 that the potential on out output 6 will follow that on line 13 as determined by the ratio between the channel conductances of the transistors 2 and 10. While the voltage at output 6 is decreasing, the voltage at output 7 increases, resulting in transistor 3 transitioning from the low Vgs region 19 of Figure 2 to the low Vds region 20. Transistor 5 similarly transitions from the low Vds region to the low Vgs region. These transitions between the regions 19 and 20 are not abrupt. However, they go via the intermediate region 18 of Figure 2 because, as described above in the context of a read operation, it is arranged that the voltage on the output 6 cannot exceed (T-0.3) volts for the transistor 5 while transistor 5 is conductive, and the voltage on the output 7 cannot exceed (T-0.3) volts for the transistor 3 while transistor 3 is conductive. A plurality of cells as described with reference to Figures 1 and 2 may be arranged in rows and columns to form a matrix of cells, individual ones or groups of which may be selected at will and read or written in response to receipt of an appropriate address and control signal. Figure 3 shows such an arrangement of n rows and m columns. Each row has a single cell access signal supply conductor 12.1 ... 12.n respectively and each column has a single pair of information signal conductors 13.1, 14.1 ... 13m, 14m respectively. The cell access signal supply conductors are fed from respective outputs A1 ... An of an address decoder 21 via a respective (electronic) switch 22.1 ... 22.n. The power supply conductors 15 and 16 of each cell 1.11 ... 1.mn are fed with 5 volts from appropriate outputs of a power supply 23. The conductors of each pair of information signal conductors 13, 14 are connected to respective inputs of a respective double changeover switch 24.1 ... 24.m. In one position of this switch the conductors are connected to a 4 volts output of supply 23, whereas in the other position they are connected via a respective double switch 25.1 .. 25.m to respective differential outputs of an input buffer 26 and also to respective differential inputs of an output buffer 27. Buffers 26 and 27 are powered with 4 volts and 5 volts respectively from the supply 23. A controller 28 has outputs C, D, E and F which are connected to contol inputs of the switches 24, control inputs of the switches 22, an output enable input of buffer 26 and an output enable input of buffer 27 respectively. In the quiescent state the switches are in the positions shown. When a cell access operation is required an appropriate address is supplied to an input 29 of decoder 21, in response to which the decoder generates +5 volts on a selected one of its outputs A1 ... An and a selected one of its outputs B1 ... Bn, the latter closing the corresponding double switch 25. The remaining outputs A and B remain at zero volts. If the required operation is a read operation a signal is also applied to a read control input 30 of controller 28. In response controller 28 first generates a pulse on its output C, changing over the switches 24, and shortly afterwards, while this pulse is still present, generates concurrent pulses on its outputs D and F, closing the switches 22 and enabling the output buffer 27. If on the other hand the required operation is a write operation a signal is supplied to a write control input 31 of controller 28. In response controller 28 first generates a pulse on its output C, changing over the switches 24. Shortly afterwards, and while the pulse on the output C is still present, it generates a pulse on its output D, closing the switches 22. Shortly afterwards, and while the pulses on the outputs C and D are still present, it generates a pulse on its output E, enabling the output of the input buffer 26. It will be noted that, because power is supplied to input buffer 26 at a level of 4 volts, the voltages supplied to the information signal conductors 13, 14 by this buffer cannot exceed this level. It will be evident that the limit chosen for the maximum positive voltages applied to the information signal conductors 13, 14 will be dependent upon the characteristics of the n-channel transistors employed in the memory cells, this limit in turn determining, together with the threshold voltages of the n-channel transistors 3 and 5 included in the cell CMOS amplifiers, the minimum ratio which the sizes of these transistors 3 and 5 can have to the sizes of the associated transistors 10 and 11.

Claims: A memory comprising a CMOS submicron Static Random Access Memory (SRAM) cell (1), said cell comprising a pair of cross-coupled inverters (2, 3, 4, 5), each inverter comprising a series connection of an inverter-PMOS-transistor (2, 4), and an inverter-NMOS-transistor (3, 5) between two supply terminals (15, 16), a supply voltage being applied across said two supply terminals in operation, outputs of the inverters being coupled to respective bit lines (13, 14) via respective NMOS-access-transistors (10, 11), the memory further comprising precharging means (24) for precharging said bit lines to a predetermined precharging voltage before executing a read operation, characterized in that said supply voltage is sufficient, when applied directly across the channel of each individual inverter-NMOS-transistor and NMOS-access-transistor while that transistor is conductive, to result in hot-electron stress in that transistor, said precharging voltage is sufficiently below said supply voltage so as to prevent hot electron stress from occurring in the access-transistors, and the inverter-NMOS-transistor (3, 5) of each of the inverters and its associated NMOS-access-transistor (10, 11) having dimensions such as to keep a voltage representing a low logical level, when such a voltage present on a respective output (6, 7) of one inverter is transmitted to the corresponding bit line, below a safety value that is sufficiently below a threshold voltage of the inverter-NMOS-transistor (5, 3) of the other inverter to prevent hot electron stress from occurring in said inverter-NMOS-transistor of the other inverter. A memory as claimed in Claim 1, characterised in that said supply voltage is substantially 5 Volts, said precharging voltage lies between approximately 2.5 Volts and 4 Volts, and said safety value is equal to approximately said threshold voltage minus 0.3 Volts. A memory as claimed in Claim 1 or 2, characterised in that the memory comprises control means for executing each write operation in such manner that it is preceded by precharging. A memory as claimed in Claim 1 or 2, characterised in that the memory comprises control means for executing each write operation in such manner that it is preceded by a read operation. A memory as claimed in Claim 1 or 2, characterised in that the width over length ratio (W/L) of each inverter-NMOS-transistor is more than approximately twice the width over length ratio (W/L) of the access-transistor which has its channel connected to the channel of that inverter-NMOS-transistor. A memory as claimed in Claim 1 or 2, characterised in that said threshold voltage is higher than approximately 1.3 Volts. A memory according to any one of the Claims 1 to 6, characterized in that, for executing a write operation, the memory comprises potential impressing and changing means arranged to change the potential on a gate electrode of each of the access transistors of the cell in a positive direction to switch these access transistors in a conductive state, prior to impressing respective potentials, one of which is more positive than the other, on the bit lines, and in that the memory includes means for charging the bit lines to a given potential prior to the change in potential on the gate electrode of each of the access transistors of the cell, and in that the more positive of said respective potentials is sufficiently less than said supply voltage to prevent substantial hot-electron stress from occurring in said access transistors.

Inventors: HARTGRING CORNELIS DIETWIN C O, POORTER TIEMEN C O PHILIPS INT, HARTGRING, CORNELIS DIETWIN C/O PHILIPS INT. B.V., POORTER, TIEMEN C/O PHILIPS INT. B.V.
Applicants: PHILIPS NV, N.V. PHILIPS' GLOEILAMPENFABRIEKEN
Application year: 1988
Main classification: G11C11
IPC codes: G11C11