BALANCED VOLTAGE-CURRENT CONVERTER AND DOUBLE-BALANCED MIXER CIRCUIT COMPRISING SUCH A CONVERTER

Abstract: The collectors of the two transistors T₅ and T₆ constitute a first and a second collector terminal to which a load circuit can be connected. The base signal currents of T₅ and T₆ are added to their collector signal currents or subtracted from their emitter signal currents by means of a first difference pair T₇, T₈ and a second difference pair T₉, T₁₀. The signal current in the first and the second collector terminal is thus very accurately equal to the input signal current at the emitters of T₅ and T₆. Due to the high voltage gain of the first and the second difference pair, T₅ and T₆ are substantially completely turned on. A double-balanced mixer circuit having a large dynamic range is provided by means of a load circuit comprising two cross-­ coupled differential amplifiers T₁, T₂ and T₃, T₄.

Description: The invention relates to a balanced voltage-current converter, comprising a first and a second transistor whose collectors constitute a first and a second collector terminal, respectively, for a load circuit connected to a common first power supply terminal and whose emitters are each connected to a common second power supply terminal by means of a current supply element, said emitters constituting inputs of the voltage-current converter. The invention also relates to a double-balanced mixer circuit comprising such a voltage-current converter. A voltage-current converter of this type having a load circuit in the form of a mixer circuit connected thereto is known from European patent application EP-Al-0131337 and is used, inter alia, in radio and television receivers and in receivers for mobile communication. The radio frequency (RF) signal from, for exmaple, an antenna is applied in phase opposition to the input terminals of the first and second transistors of the voltage-current converter and converted thereby into a proportional current variation which is superimposed via the first and second collector terminals, respectively on the emitter currents of the transistors of two difference pairs connected thereto. A mixer signal applied to these difference pairs switches their transistors so that signals having the sum of and the difference between the frequencies of the mixer signal and the radio-frequency signal are present on the collectors of the difference pairs. The signal having the difference between the frequencies, which is an intermediate frequency (IF) signal, can be taken off by means of a bandpass or lowpass filter for further processing. In such a mixer circuit non-linearities produce intermodulation products and noise which directly influence the dynamic range of the receiver. The dynamic range which is free from intermodulation is defined as the difference, expressed in dB, between the minimum signal power required for noticeable reception and the maximum signal power for which reception which is not interfered with by intermodulation products is possible. It has been found that the quality of the voltage-current converter mainly determines the quality of the mixer circuit, that is to say, the dynamic range. In the mixer circuit known from the above-mentioned European patent application EP-Al-0131337 the first and second transistors of the voltage-current converter are arranged in a common base configuration. The noise and distortion produced by the mixer circuit are largely determined by these transistors and the impedance of the signal source arranged between the input terminals of the voltage-current converter. The dynamic range of the mixer circuit can be increased by increasing the bias current of these transistors so that the ratio between the impedance of the RF signal source and the impedance of these transistors, viewed at the input terminals of the voltage-current converter, is increased so that these transistors are driven to a relatively small extent. However, this solution has the drawback that the impedance of these transistors is not constant, which also causes distortion. Moreover, the bias current in integrated circuits cannot be increased without problems because it is usually limited to a predetermined value in connection with dissipation requirements. It is therefore an object of the invention to provide a voltage-current converter which as a larger linear range (larger dynamic range) so that, for example, an improvement of both the intermodulation and the noise behaviour is obtained when it is used, for example, in a double-balanced mixer circuit of the type set forth in the opening paragraph. According to the invention this is achieved in that the voltage-current converter also comprises a first difference pair comprising a third and a fourth transistor whose emitters are connected to a common first emitter terminal, a second difference pair comprising a fifth and a sixth transistor whose emitters are connected to a common second emitter terminal, said first and second emitter terminals each being connected to the second power supply terminal by means of a current supply element, the bases of the fourth and the fifth transistor being coupled to the emitters of the first and the second transistor respectively, the collectors of the third and the sixth transistor being coupled to the first and second collector terminals, respectively, or to the emitters of the first and the second transistors, respectively, and the collectors of the fourth and the fifth transistor being coupled to the bases of the first and the second transistor, respectively, each collector being connected to the first power supply terminal by means of a current supply element. The invention is based on the recognition that the portion of the input signal current flowing in the bases of the first and the second transistor is added to the respective collectors of the first and the second transistor by means of the first and the second difference pair, respectively, or that it is subtracted from the emitter currents thereof. It is achieved thereby that the signal current in the first and second collector terminals is very accurately equal to the input signal current of the first and the second transistor, respectively. The logarithmic base-emitter voltage of these transistors is supplied by the fourth and the fifth transistor, respectively, each of which has a high voltage gain. This means that the first and the second transistor can be almost fully turned on so that the dynamic range of the balanced voltage-current converter according to the invention is larger than that of the converter as is used in the known double-balanced mixer circuit. Other differential amplifiers with improved linear amplification are known for example from US-A-4536717 and EP-A-0090543 but they use a different approach for achieving this result. The noise produced by the circuit according to the invention is mainly determined by the transistors of the first and the second difference pair. In a preferred embodiment of the invention the noise contribution of these difference pairs is reduced in that the bases of the third and the sixth transistor are at a fixed potential as far as signal currents are concerned, a seventh and an eighth transistor having their emitters connected to ground with respect to the signal current, the bases of the third and the sixth transistor being coupled to the collectors of the seventh and the eighth transistor, respectively, which collectors being connected to the first power supply terminal by means of a current supply element and the bases of the seventh and the eighth transistor being connected to the bases of the fourth and the fifth transistor, respectively. The operation of the seventh and the eight transistor may essentially be compared with that of broadband operational amplifiers whose bases constitute the inverting inputs and whose emitters constitute the non-inverting inputs,which amplifiers are negatively fed back via the first and the second difference pair, respectively. This preferred embodiment of the invention has the advantage that the noise produced by the circuit is now mainly determined by the seventh and the eighth transistor which have a relatively low bias current so that their current noise contribution is small. As a result of the negative feedback the input impedance of the voltage-current converter is further reduced in an electronic manner. Since, as has already been stated, the fourth and the fifth transistor have a high voltage gain, it is desirable to make this gain narrow-band so as to prevent unstable behaviour of the voltage-current converter according to the invention. In a further embodiment of the invention this is achieved in that elements having a capacitive action are arranged between the base of the first transistor and the first emitter terminal and between the base of the second transistor and the second emitter terminal, respectively. This arrangement does not yield any extra distortion, because the considerably non-linear currents through these elements having a capacitive action flow in a closed loop through the fourth and the fifth transistor, respectively. The currents flowing in the first and second collector terminals remain equal to the respective input currents of the voltage-current converter. To prevent further instabilities in the circuit according to the invention, the bandwidth of the respective feedback loops constituted by the first and the second difference pair should be limited. In yet another embodiment of the invention this is achieved in that the collectors of the seventh and the eighth transistor are each connected to ground by means of further elements having a capacitive action. In yet another embodiment of the invention the less favourable high-frequency properties of the third and the sixth transistor arranged as emitter followers can be improved in that further elements having a capacitive action are arranged between the base of the third transistor and the first emitter terminal and between the base of the sixth transistor and the second emitter terminal, respectively. As already stated, the input signal is to be applied in phase opposition to the emitters of the first and the second transistor. If the signal source is designed has a floating source, it may be directly connected between the emitters. If one pole of the signal voltage source carries a fixed voltage, because it is connected, for example, to ground, the input voltage can be applied in phase opposition to the emitters in known manner by means of a transformer having one primary winding and two symmetrical secondary windings. If the use of a transformer is difficult or undesirable, for example from a circuit-technical point of view, this drawback can be obviated in yet another embodiment of the invention in that the converter further comprises a third and a fourth difference pair comprising a seventh and a ninth and an eighth and a tenth transistor, respectively, the emitters of each difference pair being connected together and to the second power supply terminal via a current supply element, the bases of the third and the sixth transistor being connected to the collectors of the seventh and the eighth transistor, respectively, which collectors are connected to the first power supply terminal by means of a current supply element, the bases of the seventh and the eighth transistor being connected to the bases of the fourth and the fifth transistor, respectively, the collectors of the ninth and the tenth transistor being coupled to the first power supply terminal with respect to the signal current and the bases of the ninth and the tenth transistor constituting the input terminals of the voltage-current converter. The circuit thus formed has a high input impedance and by connecting one of the input terminals to ground a non-floating signal source can be connected in a simple manner without the use of a transformer. This embodiment of the invention can be manufactured as one assembly in an integrated manner. As described in the foregoing, a voltage-current converter according to the invention may advantageously be used in a double-balanced mixer circuit comprising a difference pair comprising an eleventh and a twelfth transistor whose emitters are connected together and which constitute a common emitter terminal, and a further difference pair comprising a thirteenth and a fourteenth transistor whose emitters are connected together and which constitute a common further emitter terminal, the collectors of the eleventh and the thirteenth transistor being connected together and constituting a common collector terminal, the collectors of the twelfth and the fourteenth transistor being connected together and constituting a common further collector terminal, each collector terminal being connected to the common first power supply terminal by means of an impedance, the bases of the eleventh and the fourteenth transistor being connected to a common base terminal, the bases of the twelfth and the thirteenth transistor being connected to a common further base terminal, between which base terminals a mixer signal can be applied, and the first and second collector terminals of the voltage-current converter being connected to the respective emitter terminals of the mixer circuit. The invention will now be described in greater detail, by way of example, with reference to the accompanying drawings in which Fig. 1 is a basic circuit diagram of a double-balanced mixer circuit in which the input signal is applied to the bases of the transistors of the voltage-current converter; Fig. 2 is a basic circuit diagram of a double-balanced mixer circuit in which the transistors of the voltage-current converter are arranged in a common base configuration and in which the input signal is applied to their emitters via a transformer; Fig. 3 is a basic circuit diagram of a double-balanced mixer circuit comprising a voltage-current converter according to the invention; Fig. 4 is a basic circuit diagram of a double-balanced mixer circuit with a preferred voltage-current converter according to the invention; Fig. 5 is a basic circuit diagram of a double-balanced mixer circuit with a further voltage-current converter according to the invention; and Fig. 6 is a circuit diagram of a practical version of the circuit of Fig. 4 in which capacitors are incorporated in the voltage-current converter to prevent unstable behaviour of the mixer circuit. The double-balanced mixer circuit shown in Fig. 1 comprises a difference pair which is constituted by the transistors T₁ and T₂ having a common first emitter terminal 3, and a further difference pair constituted by the transistors T₃ and T₄ having a common further emitter terminal 4. The bases of the transistors T₁ and T₄ are connected to a common base terminal 7, whilst the bases of the transistors T₂ and T₃ are connected to a common further base terminal 8. The collectors of the transistors T₁ and T₃ are connected to a common collector terminal 5, whilst the collectors of the transistors T₂ and T₄ are connected to a common further collector terminal 6. The collector terminal 5 is connected by means of an impedance 9 to a common first power supply terminal 1 and the further collector terminal 6 is also connected to the first power supply terminal 1 by means of an impedance 10. The impedances 9 and 10 may be, for example resistors. An oscillator signal v osc by means of which the transistors of the first and the second difference pair are switched is applied to the base terminals 7 and 8. The voltage-current converter is constituted by the transistors T₅ and T₆. The RF input signal v in from, for example, an antenna is applied to input terminals 14 and 15 which are connected to the bases of the transistors T₅ and T₆, respectively, whose collectors are connected to the emitter terminal 3 and to the further emitter terminal 4, respectively. The emitters of the transistors T₅ and T₆ are each connected, via current supply elements 11 and 12 respectively, to a common second power supply terminal 2. An impedance 13, for example in the form of a resistor, is arranged between the emitters of the transistors T₅ and T₆. The first power supply terminal 1 is connected to a positive pole of a voltage source and the second power supply terminal 2 is connected to a negative pole. The oscillator signal and the RF signal are mixed by means of the mixer circuit so that signals having the sum of and the difference between the frequencies of the oscillator signal and the RF signal appear at the collector terminals 5, 6. The signal having the difference between the frequencies, the IF signal, can be taken off by means of a bandpass or lowpass filter. The transistors T₅ and T₆ constitute a so-called voltage-current converter in which the signal voltage v in applied to the input terminals 14, 15 is converted into a proportional signal current i in through the impedance 13, which signal current flows via the emitter terminal 3 and the further emitter terminal 4 because the current supply elements 11, 12 have an infinitely high impedance in the ideal case. In this Application the term current supply element is understood to mean a circuit through which a given substantially constant direct current flows and whose impedance for alternating current is relatively large. Although in the circuit of Fig. 1 current sources are shown, the envisaged action can be alternatively realised, for example, by resistors of suitable value. A mixer circuit of the type described above has the advantage that a satisfactory separation is obtained between the input signal, the output signal and the oscillator signal, that little oscillator power is required and that the circuit is integrable. Properties such as noise and distortion are largely determined by the transistors T₅, T₆ and the resistor 13. Known measures for improving these properties comprise the split-up of the conversion resistor 13 into a series arrangement of two resistors of half the value, whilst the two current supply elements 11, 12 are replaced by one current supply element of double the value connected between the junction point of the two resistors of half value and the second power supply terminal 2. It has been found that the linearity and hence the intermodulation behaviour as well as the noise behaviour can be improved by modifying the mode of driving the transistors T₅ and T6 of the voltage-current converter, as is shown in Fig. 2. In this Figure the bases of the transistors T₅ and T₆ are connected to ground with respect to the signal current, whilst the input signal is presented between their emitters i.e. the input terminals 19 and 20, respectively. The dynamic range of the total mixer circuit in this embodiment is larger than in the circuit of Fig. 1 because this is now substantially entirely determined by the dynamic range of the common base configuration thus formed. If the RF signal originates from a so-called floating source, it can be connected in series with a conversion resistor to the input terminals 19, 20. Since the RF input signal usually comes from a voltage source 17 having a source impedance 18, with one pole of the source connected to ground, a floating source has to be created to drive the emitters of the transistors T₅ and T6. In Fig. 2 this has been realised by means of a symmetrical transformer 16 which converts the RF input signal into two signals having opposite phases. The transformer has a primary winding 16a one side of which is connected to ground and to the other side of which the RF signal source 17 is connected via source impedance 18. The transformer also comprises two secondary windings 16b and 16c having a common connection point which, as is shown, may be connected to ground and the other connection points of which are connected to the input terminals 19 and 20, respectively. For obtaining optimum noise and intermodulation behaviour, the linearity of the circuit can be adjusted by means of the ratio between the input impedance of the common base configuration and the source impedance of the signal source. The value of the input impedance of the common base configuration is dependent on the value of the bias current I₀ of the two current supply elements 11, 12. The value of the source impedance between the input terminals 19, 20 can be adjusted by means of the transformation ratio of the transformer 16. As already stated hereinbefore, the ratio between the transformed source impedance and the input impedance of the cmmon base configuration should be chosen to be as large as possible in order to obtain a large linearity. As already noted in the opening paragraph this circuit has the drawback that the input impedance of the common base configuration is not constant, which causes distortion. Moreover, by increasing the bias current of the transistors T₅ and T₆, the current noise contribution thereof is increased. A already stated in the opening paragraph, the value of the bias currents is limited in integrated circuits in connection with the heat dissipation of the entire circuit. Fig. 3 shows the basic circuit diagram of an example of a double-balanced mixer circuit comprising a voltage-current converter according to the invention. In addition to the known transistors T₅ and T₆ described with reference to Fig. 2 the voltage-current converter also comprises two difference pairs constituted by the transistors T₇, T₈ and T₉, T₁₀, respectively. The emitters of the transistors T₇ and T₈ of the first difference pair are connected to a common first emitter terminal 21 and the emitters of the transistors T₉ and T₁₀ of the second difference pair are connected to a common second emitter terminal 22. The bases of the transistors T₇ and T₁₀ are connected to ground with respect to the signal current. The collectors of the transistors T₅ and T₆ constitute a first and a second collector terminal, respectively, to which the collectors of the transistors T₇ and T₁₀ and the emitter terminal 3 and the further emitter terminal 4 of the circuit constituted by T1, T2, T₃ and T₄ are connected. The collector of transistor T₈ is connected to the base of transistor T₅ and to the first power supply terminal 1 via a current supply element 25. The collector of transistor T₉ is connected to the base of transistor T₆ and also to the first power supply terminal 1 via a current supply element 26. The base of transistor T₈ is connected with respect to the signal current to the emitter of transistor T₅ and the base of transistor T₉ is connected with respect to the signal current to the emitter of transistor T₆. The first emitter terminal 21 and the second emitter terminal 22 are each connected by means of a current supply element, 23 and 24 respectively, to the second power supply terminal 2. An input signal v in can be applied in the manner described to the input terminals 19, 20. The operation of the voltage-current converter according to the invention will hereinafter be described with reference to the portion of the circuit connected to the emitter terminal 3. Based on the symmetrical structure of the circuit the portion connected to the further emitter terminal 4 operates in a corresponding manner. The logarithmic base-emitter voltage of transistor T₅ is supplied by transistor T₈ of the first difference pair. Assuming that the current supply element 25 is ideal, the signal current flowing in the collector lead of transistor T₈ is equal to the signal current in the base lead of transistor T₅. Assuming that the current supply element 23 is also ideal, the signal current flowing in the collector lead of transistor T₇ is substantially equal and opposite to the collector current of transistor T₈, thus in this case it is equal to the base current of transistor T₅. Since the collector of transistor T₇ is connected to the emitter terminal 3, the total signal current flowing in this emitter terminal 3 is equal to the signal current in the emitter lead of transistor T₅. If the current flowing in the base leads of the transistors T₇ and T₈ is ignored, the input current of the circuit in the input terminal 19 is thus very accurately equal to the signal current in the collector terminal of the voltage-current converter connected to the emitter terminal 3. Instead of connecting the collector of transistor T₇ to the first collector terminal of the voltage-current converter, it may alternatively be connected to the emitter of transistor T₅ as is shown by means of a broken line in Fig. 3. The signal current flowing in the collector lead of transistor T₇, which current is equal to the base current of transistor T₅ as far as its value is concerned, is then subtracted from the emitter current of T₅. If the signal currents in the base leads of the transistors T₇ and T₈ are ignored, it now also holds that the input current of the circuit in the input terminal 19 is very accurately equal to the signal current in the first collector terminal. Of course, in the balanced circuit the collector of transistor T₁₀ should be connected in a corresponding manner to the emitter of transistor T₆. The input impedance of the voltage-current converter between the input terminals 19, 20 is now much smaller due to the considerable negative feedback around the transistors T₅ and T₆ so that the distortion resulting from the variations in the input impedance of the circuit is reduced to a very considerable extent. Since the logarithmic base-emitter voltages of the transistors T₅ and t₆ are supplied by the transistors T₈ and T₉, respectively, which latter transistors have a high voltage gain, the transistors T₅ and T₆ can be substantially fully turned on so that the dynamic range of the mixer circuit is very large. The noise produced by the circuit is mainly determined by the transistors of the first and the second difference pair. In the preferred embodiment of the invention shown in Fig. 4 further negative feedback is provided by means of the transistors T₁₁ and T₁₂ as compared with the embodiment of Fig. 3. The collector of transistor T₁₁ is connected to the base of transistor T₇ and to the first power supply terminal 1 via a current supply element 27. The emitter of transistor T₁₁ is connected to ground with respect to the signal current and its base is connected to the base of transistor T₈. The collector of transistor T₁₂ is connected to the base of transistor T₁₀ and to the first power supply terminal 1 via a current supply element 28. The emitter of transistor T₁₂ is also connected to ground with respect to the signal current and its base is connected to the base of transistor T₉. The transistors T₁₁ and T₁₂ arranged in this manner essentially function as a broadband operational amplifier in which the bases constitute the inverting inputs, the emitters constitute the non-inverting inputs and the collectors constitute the outputs. The operational amplifiers formed in this manner are negatively fed back via the first and second difference pairs, respectively. The input impedance of the mixer circuit between the input terminals 19, 20 is further reduced with respect to the circuit of Fig. 3, whilst the noise is mainly determined by the transistors T₁₁ and T₁₂ which moreover have a relatively low bias current and consequently a low current noise contribution. However, if a stable high input impedance is necessary, the transistors T₁₁ and T₁₂ can easily be formed as components of difference pairs, for example, as is shown in Fig. 5. The emitter of transistor T₁₁ is connected to the emitter of a transistor T₁₃ and is connected to the second power supply terminal 2 via a current supply element 29. Similarly, the emitter of transistor T₁₂ is connected to the emitter of a transistor T₁₄ and to the second power supply terminal 2 via a current supply element 30. The collectors of the transistors T₁₃ and T₁₄ may be connected, as is shown, with respect to the signal current to the first power supply terminal 1. The bases of the transistors T₁₃ and T₁₄ now constitute the high-ohmic input terminals 31, 32 of the voltage-current converter, an impedance 13 being arranged between the terminals 19 and 20 in conformity with the circuit according to Fig. 1. Otherwise, the circuit operates in the same way as described hereinbefore with reference to Fig. 4 and it can be completely integrated. In the basic circuit diagram of the preferred voltage-current converter according to the invention shown in Fig. 4 two gain loops can be distinguished in each of the branches connected to the emitter terminal 3 and the further emitter terminal 4, respectively, which might give rise to unstable behaviour of the mixer circuit. Fig. 6 shows a practical version of the circuit of Fig. 4 in which capacitors are incorporated at various points to prevent instabilities in the circuit. The current supply elements 25, 26, 27 and 28 are constituted by the transistors Q₂₅, Q₂₆, Q₂₇ and Q₂₈ with their respective resistors R₂₅, R₂₆, R₂₇ and R₂₈ and the diodesD₂₅, D₂₇, resistor R₃₃ and the diodes D₂₆, D₂₈ and resistor R₃₄ arranged in known manner as is shown in Fig. 6. The current supply elements 11, 12, 23 and 24 are constituted by the resistors R₁₁, R₁₂, R₂₃ and R₂₄, respectively. Series arrangements of two diodes D₁₁, D₁₃ and D₁₂, D₁₄ are incorporated in the emitter leads of the transistors T₁₁ and T₁₂, respectively. The connection between the emitter of transistor T₅ and the base of transistor T₈ incorporates a resistor R₅ and the connection between the emitter of transistor T₆ and the base of transistor T₉ incorporates a resistor R₆. As has been described with reference to Fig. 2, the RF input signal is applied to the input terminals 19, 20 by means of a symmetrical transformer 16. As described in the foregoing, the transistors T₁₁ and T₁₂ essentially constitute broadband operational amplifiers which are negatively fed back by means of the third and the fourth difference pair, respectively. To maintain these feedback loops stable, the collector of transistor T₁₁ is connected to ground by means of a capacitor C₁₁ and the collector of transistor T₁₂ is connected to ground by means of a capacitor C₁₂. The total bandwidth is laid down by these capacitors and the effective transductance of the transistors T₁₁ and T₁₂. The parasitic capacitances which are always present in the circuit already have some stabilising effect corresponding to the influence of C₁₁ and C₁₂. Essentially, three frequency-dependent impedances can be distinguished at the bases of each of the transistors T₅ and T₆, namely the collector impedance of the transistors T₈, T₉, the impedance of the current supply elements 25, 26 and the input impedance of the transistors T₅ and T₆ themselves. Together with the parasitic capacitances which are present, these impedances give rise overall to a relatively large time constant which produces a first corner frequency f₁ in the frequency characteristic of the open loop gain. The time constant of the transistors T₅, T₆ themselves produces a second corner frequency f₂ in the frequency characteristic in which f₂ is larger than f₁. In the case of full negative feedback, that is to say transistor T₅ drives transistor T₈ and transistor T₆ drives transistor T₉, a phase shift of 180 o at unity gain may occur at the frequency f₂. This loop can be rendered stable by increasing the first time constant so that the dominant corner frequency f₁ shifts to a lower value and the gain at the second corner frequency f₂ becomes smaller than one so that the loop becomes stable. An increase in the time constant at the bases of the transistors T₅ and T₆ is realised by means of capacitors C₅, C₆ arranged between these bases and the first emitter terminal 21 and the second emitter terminal 22, respectively. In principle, these capacitors may alternatively be arranged between the bases of the transistors T₅ and T₆ and ground. However, the currents flowing through the respective capacitors, which currents are not linear due to the base-emitter voltages, are then added to the collector currents of the respective transistors T₅ and T₆ via T₇, T₈ and T₉, T₁₀, respectively, but they do not flow in their emitter leads, which may give rise to distortion. By connecting the relevant capacitors C₅ and C₆ to the first emitter terminal 21 and the second emitter terminal 22, respectively, the currents therethrough flow in closed loops constituted by the capacitor C₅ and the transistor T₈, and the capacitor C₆ and the transistor T₉, respectively. The signal currents flowing in the collector terminals of the voltage-current converter then remain equal, according to the invention, to the respective signal currents in the emitter leads of the respective transistors T₅ and T₆. The relatively poor RF properties of the transistors T₇ and T₁₀ arranged as emitter followers may be improved by capacitors C₇ and C₁₀ arranged between their respective bases and emitters. For RF signals the capacitors C₇ and C₁₀ constitute so-called "feed forward" capacitances in the feedback loops constituted by C₇, C₅, T₅ and C₁₀, C₆, T₆, respectively. The symmetrical transformer 16 used in Fig. 6 for giving the signal source having one side grounded a floating nature can be replaced by a differential amplifier having a so-called differential output. The diodes D₁₁, D₁₃ and D₁₂, D₁₄ may alternatively be replaced by a single diode screen pair or may be decoupled for signal currents. The use of the transformer shown in Fig. 6 for a non-floating signal source can be avoided by connecting one of the input terminals 31 or 32 to ground in the circuit according to Fig. 5 and connecting the signal source 17 having one side grounded to the other, free input terminal. In the embodiment of the invention shown all transistors are NPN transistors. Instead, PNP transistors may of course be used which transistors, however, have a poorer RF behaviour than NPN transistors in an integrated circuit. It is of course also possible to use field effect transistors instead of bipolar transistors and to replace the terms emitter, collector and base in this Application by the terms source, drain and gate, respectively. It is also possible to incorporate resistors and capacitors for decoupling at various locations in the circuit or to obtain certain other desired properties, all this within the scope of the invention as defined by the claims. Although the voltage-current converter according to the invention has been described hereinbefore with reference to its use in a double-balanced mixer cirucit, it is of course not limited thereto as far as its fields of use are concerned. Other fields of use are, for example, automatic gain control circuits, impedance converters, etc.

Claims: A balanced voltage-current converter comprising a first and a second transistor (T5, T6) whose collectors constitute a first and a second collector terminal, respectively, for a load circuit (3-10) connected to a common first power supply terminal (1) and whose emitters are each coupled to a common second power supply terminal (2) by means of a current supply element (11, 12), said emitters constituting inputs of the voltage-current converter, characterized in that the voltage-current converter also comprises a first difference pair comprising a third (T7) and a fourth (T8) transistor whose emitters are connected to a common first emitter terminal, a second difference pair comprising a fifth (T9) and sixth (T10) transistor whose emitters are connected to a common second emitter terminal, said first and second emitter terminals each being connected to the second power supply terminal by means of a current supply element (23, 24), the bases of the fourth and the fifth transistor being coupled to the emitters of the first and the second transistor, respectively, the collectors of the third and the sixth transistor being coupled to the first and second collector terminals, respectively, or to the emitters of the first and the second transistor, respectively, and the collectors of the fourth and the fifth transistor being coupled to the bases of the first and the second transistor, respectively, each collector being connected to the first power supply terminal by means of a current supply element (25, 26). A balanced voltage-current converter as claimed in Claim 1, characterized in that the bases of the third (T7) and the sixth (T10) transistor are at a fixed potential as far as signal currents are concerned, in that a seventh (T11) and an eighth (T12) transistor have their emitters coupled to ground, the bases of the third and the sixth transistor being connected to the collectors of the seventh and the eighth transistor, respectively, which collectors being connected to the first power supply terminal by means of a current supply element (27, 28) and the bases of the seventh and the eighth transistor being connected to the bases of the fourth (T8) and the fifth (T9) transistor, respectively. A balanced voltage-current converter as claimed in Claim 1, characterized in that the emitters of the first (T5) and the second (T6) transistor are connected together by means of an impedance (13), and in that the converter further comprises a third and a fourth difference pair, which difference pairs comprise a seventh (T11) and a ninth (T13) and an eighth (T12) and a tenth (T13) transistor, respectively, the emitters of each difference pair being connected together and to the second power supply terminal via a current supply element (29, 30), the bases of the third (T7) and the sixth (T10) transistor being connected to the collectors of the seventh (T11) and the eighth (T12) transistor, which collectors are connected to the first power supply terminal by means of a current supply element (27, 28), the bases of the seventh and the eighth transistor being connected to the bases of the fourth (T8) and the fifth (T9) transistor, respectively, the collectors of the ninth and the tenth transistor being coupled to the first power supply terminal and the bases of the ninth and the tenth transistor constituting input terminals of the voltage-current converter. A balanced voltage-current converter as claimed in Claim 1, 2 or 3, characterized in that elements having a capacitive action (C5, C6) are arranged between the base of the first transistor (T5) and the first emitter terminal and between the base of the second transistor (T6) and the second emitter terminal, respectively. A balanced voltage-current converter as claimed in Claim 2, 3 or 4 when appendant to Claims 2 and 3, characterized in that the collectors of the seventh (T11) and the eighth (T12) transistor are each connected to ground by means of further elements (D11-D14) having a capacitive action. A balanced voltage-current converter as claimed in Claim 1, 2, 3, 4 or 5, characterized in that further elements (C7, C10 having a capacitive action are arranged between the base of the third transistor (T7) and the first emitter terminal and between the base of the sixth (T10) transistor and the second emitter terminal, respectively. A double-balanced mixer circuit comprising a voltage-current converter as claimed in any one of the preceding Claims, comprising a difference pair comprising an eleventh (T1) and a twelfth (T2) transistor whose emitters are connected together and which constitute a common emitter terminal, and a further difference pair comprising a thirteenth (T3) and a fourteenth (T4) transistor whose emitters are connected together and which constitute a common further emitter terminal, the collectors of the eleventh and the thirteenth transistor being connected together and constituting a common collector terminal, the collectors of the twelfth and the fourteenth transistor being connected together and constituting a common further collector terminal, each collector terminal being connected to the common first power supply terminal by means of an impedance, the bases of the eleventh and the fourteenth transistor being connected to a common base terminal, the bases of the twelfth and the thirteenth transistor being connected to a common further base terminal, between which base terminals a mixer signal can be applied, and the first and second collector terminals of the voltage-current converter being connected to the respective emitter terminals of the mixer circuit.

Inventors: SEMPEL, ADRIANUS
Applicants: N.V. PHILIPS' GLOEILAMPENFABRIEKEN
Application year: 1988
Main classification: H03D7
IPC codes: H03F3, H03D7