Microwave discharge light source apparatus

Abstract: A power supply circuit for a magnetron adapted to supply microwave energy to an electrodeless discharge bulb is disclosed the circuit comprises a rectifier coupled across a commercial AC voltage source, a filter for smoothing the output of the rectifier, an inverter for converting the DC voltage supplied from the filter into a high frequency AC voltage, a step-up transformer for stepping up the high frequency AC voltage outputted from the inverter, and a rectifier which rectifies the high voltage AC output of the transformer into a unidirectional voltage which is supplied to the magnetron. The inverter switching is controlled by a pulse width modulation control circuit to maintain the magnetron output power at a predetermined level. According to one aspect, an inductance is provided in the circuit which supresses high frequency components in the currents flowing through the windings of the transformer; according to another aspect, the inverter switching frequency (expressed in kHz) is set at a value not less than 1500/D, wherein D represents the diameter of the electrodeless bulb expressed in millimeters; according to still another aspect, the peak to the mean value ratio of the magnetron current is limited under 3.75 inclusive.

Description: TECHNICAL FIELD The present invention relates to a microwave discharge light source apparatus with an electrodeless discharge bulb and a circuit system adapted to supply microwave energy to said bulb. Said circuit system includes a magnetron and a power supply circuit therefor. BACKGROUND ART In recent years, microwave discharge light source having an electrodeless bulb disposed in a microwave resonance cavity has been developed and is attracting attention because of its long life. Fig. 1a shows one of such microwave discharge light source apparatus disclosed in Japanese Laid-Open Patent Application 56-126250; Fig. 1b shows a modification thereof disclosed in Japanese Laid-Open Patent Application 57-55091. In both apparatuses, a magnetron 1 having an antenna 1a is disposed at the end of a waveguide 2 having ventilating holes 2a which supplies the microwave generated by the magnetron 1 to a resonance cavity 3 through a microwave supply port 3a; the cavity 3 is formed by a paraboloidal wall 3b having a light reflecting rotationally symmetric inner surface and a metallic mesh 3c forming the front face of the cavity 3, which opaque to microwave but transparent to light. A spherical electrodeless discharge bulb 4 disposed in the cavity 3 and having encapsulated therein a plasma generating medium emitts light through the metallic mesh 3c covering the front face of the cavity 3, when the microwave is radiated into the bulb 4: at first, the gas enclosed in the bulb 4 undergoes discharge due to the microwave radiated into the cavity 3; thus, the inner surface of the bulb 4 is heated, and the metal, such as mercury, deposited on the inner surface of the bulb 4 is evaporated into a gas; as a result, the discharge in the bulb 4 goes over to that of the metallic gas, in which light having an emission spectrum peculiar to the kind of the metal is emitted from the discharging metallic gas. The emitted light is reflected by the cavity wall 3b and is radiated forward through the front mesh 3c. The apparatuses further comprise a fan 5 at the end wall of the housing 6 for cooling the magnetron 1 and the bulb 4. Microwave discharge light source apparatuses similar to those described above are also disclosed in U.S. Patent Nos. 4,498,029 and 4,673,846, both issued to Yoshizawa et al. The first of these U.S. Patents teach an apparatus in which the bulb is sufficiently small to act substantially as a point light source; the second teach an apparatus in which the wall surface of the microwave resonance cavity having the electrodeless bulb disposed therein is mostly constituted by a mesh, wherein the wires constituting the mesh are electrically connected each other without any contact resistance. A conventional power supply circuit for a magnetron is disclosed in Japanese Laid-Open Utility Model Application 56-162899, or in the first of the above mentioned U.S. Patents, according to which a commercial voltage source at 50 to 60 Hz is coupled to a step-up transformer, and the resulting stepped-up high-voltage AC current is rectified by a full-wave rectifier circuit to obtain pulsing unidirectional current which is supplied to the magnetron. As the rectification is effected by a full-wave rectifier circuit, the resulting high voltage rectified current pulsates at 100 to 120 Hz; consequently, the magnetron generates a microwave pulsing at 100 to 120 Hz. Thus, when magnetron 1 is supplied by this conventional circuit, the discharge in the bulb 4 is caused by the microwave pulsing at 100 to 120 Hz. The disadvantage of this type of conventional power supply circuit is as follows. First, as the commercial AC voltage of relatively low frequency, i.e., 50 to 60 Hz, is directly supplied to the primary winding of the step-up transformer to obtain a high voltage needed to supply the magnetron, the transformer should be provided with a heavy iron core; the weight of the transformer is equal to or greater than 10 kg when the input power to the magnetron is 1.5 kW. Second, as a full-wave rectifier circuit is used to rectify the AC current induced in the secondary winding of the transformer, neither one of the terminals of the secondary winding can be grounded; thus, the over-all size of the transformer should be further increased to ensure an electrical insulation thereof; in addition, extremely high voltage may develop in portions within or outside of the transformer, which diminishes the reliability of the parts thereof. If the rectifier circuit coupled to the secondary winding of the transformer is constituted by a half-wave rectifier circuit, one terminal of the secondary winding of the step-up transformer can be grounded to minimize the above-mentioned drawbacks of the conventional power supply circuit. This, however, causes another problem: as the voltage applied to the magnetron 1 is reduced to 0 during the half period of the commercial AC voltage cycle, the generation of the microwave is stopped for about 8 to 10 ms; thus there is the danger that the discharge is extinguished during the same time intervals. Thus, a full-wave rectifier circuit must have been used to rectify the outputs of the step-up transformer. Fig. 2a shows an inverter type power supply circuit for a magnetron taught in Japanese Patent Publication 60-189889, wherein the magnetron 1 is supplied by the circuit as described in what follows. A rectifier circuit 8 is coupled across the lines of a commercial AC voltage source E; a pair of series-connected capacitors C1 and C2 are coupled across the output terminals of the rectifier circuit 8 to obtain a substantially constant voltage DC power. An oscillator circuit 9, which comprises a Zener diode Zn, a capacitor C3, a plurality of resistors, and an amplifier A, is coupled across the capacitor C2 to output a rectangular waveform signal having a frequency substantially higher than that of the commercial AC voltage source E to a control circuit 10 comprising a transistor T1, a diode D1, and a plurality of resistors; the frequency of the rectangular waveform signal of the oscillator circuit 9 is determined by the values of the resistors and the capacitor C3 thereof. The control circuit 10 controls the alternate switching actions of a switching circuit comprising the power transistors 11 and 12 and the controlling transistors 11a and 12a therefor. Namely, by alternately turning on and off the controlling transistors 11a and 12a, the circuit 10 alternately turns on and off the power transistors 11 and 12 in response to the output signal of the oscillator circuit 9. Thus, a high frequency rectangular waveform AC current is supplied to the primary winding P of the transformer T through a filter circuit 13. The AC voltage induced in the secondary winding S of the transformer T is rectified by a voltage doubler rectifier circuit consisting of a capacitor C4 and a diode D2, and is supplied therefrom to the magnetron 1. The inverter type power supply for a magnetron as described above also suffers disadvantages. Namely, as the magnetron 1 constitutes a non-linear load, the output power and current thereof and the inverter current supplied to the step-up transformer become unstable when the voltage level of the voltage source E fluctuates; the over-current resulting therefrom may destroy the power transistors 11 and 12. Fig. 2b shows another inverter type power supply circuit for a magnetron taught in Japanese Laid-Open Patent Application 62-113395, wherein the magnetron 1 is supplied by the circuit as follows. A diode bridge rectifier circuit 8 comprising four diodes Do is coupled across the commercial AC voltage source E; a smoothing filter circuit 9 consisting of a capacitor Co is coupled across the output terminals of the rectifier circuit 8 to output a substantially constant DC voltage therefrom. The switching circuit 10 comprises switching transistors Q1 and Q2 and diodes D1 and D2 for reverse currents coupled across the source and the drain thereof, respectively, the transistors Q1 and Q2 being coupled across the negative output terminal of the filter circuit 9 and the terminals P1 and P2 of the primary winding P of the transformer T, respectively. The positive output terminal of the filter circuit 9 is coupled to the center tap 0 of the primary winding P of the transformer T. The gate terminals g1 and g2 of the transistors Q1 and Q2, respectively, is coupled to the center tap 0 of the primary winding P of the transformer T. The gate terminals g1 and g2 of the transistors Q1 and Q2, respectively, are coupled to the output terminals of a control circuit 11. The voltage doubler rectifier circuit 12 consisting of series-connected capacitor C1 and a diode D3 is coupled across the terminals S1 and S2 of the secondary winding S of the transformer T; the negative output terminal d of the rectifier circuit 12 is coupled to the cathode K of the magnetron 1, which is heated by a filament current supplied thereto from a commercial AC voltage source through an electrically insulating transformer (not shown) and the lines h; the positive output terminal f of the rectifier circuit 12, on the other hand, is coupled to the anode A of the magnetron 1 through a resistor R, the terminals of the resistor R being coupled to the input terminals of the control circuit 11. The control circuit 11 outputs pulses to the transistors Q1 and Q2 at a varying frequency centered around a fixed frequency, to alternately turn on and off the transistors Q1 and Q2. Thus, the current flows alternately from the center tap 0 to the terminal P1 and to the terminal P2 of the primary winding P of the transformer T to induce an AC voltage in the secondary winding S thereof, which is rectified by the rectifier circuit 12 and supplied therefrom to the magnetron 1. The pulse signals of the control circuit 11 at the fixed frequency are subjected to frequency modulation utilizing a modulating signal having a frequency which is lower than the frequency of the fixed frequency of the output pulse signals, to prevent flickering of the discharge in an electrodeless bulb such as those shown in Figs. 1a and 1b; the flickering of the discharge is caused by an acoustic resonance in the bulb due to the ripple or fluctuation of the microwave energy. Further, the circuit 11 varies the length of time during which the transistors Q1 and Q2 are turned on, so that the output power of the magnetron is held constant irrespective of the fluctuation in the voltage source level; this can be effected by detecting the magnetron current by means of the voltage drop across the resistor R, thanks to the substantially constant voltage characteristic of the magnetron 1. The inverter type power supply circuit for a magnetron described just above is small-sized and is effective to a certain degree to prevent the flickering of the discharge arc of the electrodeless discharge bulb, thanks to the adoption of the high frequency inverter in the circuit. The flickering of the discharge arc, however, may persist even in the apparatuses supplied by the circuit, depending on the kind and amount of the material encapsulated in the bulb and on the microwave energy level radiated into the bulb: the flickering of the arc is particularly manifest when a metal halide compound such as sodium iodide is encapsulated in the bulb in addition to mercury and a starter rare gas, or when the microwave energy supplied to the bulb is at a high level. Further disadvantage of the circuit of Fig. 2b is that the controlling circuit 11 thereof has a complicated structure, because the pulse signals thereof are subjected to frequency modulation and the length of the turning-on time of the switching is varied to maintain the output power of the mangetron 1 at a constant level. Power supply circuits for a magnetron utilizing inverters are also disclosed in U.S Patent No. 4,593,167 issued to Nilssen and U.S. patent No. 3,973,165 issued to Hester. The first of these U.S. patents teach a power supply circuit for a magnetron of a microwave oven including an inverter, wherein the step-up transformer exhibits relatively high leakage between its input and output windings and a capacitor is connected across the step-up transformer's output winding; further, a rectifier and filter means is connected in parallel with the capacitor, and supplies substantially constant DC voltage to the magnetron. The second U.S. patent teach an inclusion of an inverter in a power supply for a magnetron which supplies microwave energy to a microwave oven, etc, wherein the DC current obtained by rectifying a commercial AC voltage of 60 Hz is supplied to the step-up transformer through an inductor, which prevents high frequency currents or voltages to flow into the AC voltage source lines. Further, Japanese Laid-Open Patent Application 62-290098 teaches a microwave discharge light source apparatus including an inverter type power supply circuit for the magnetron, wherein the inverter frequency is set at a few tens kHz, for example, thereby maintaining parameters of the plasma in the bulb at a substantially constant level to prevent the flickering of the discharge in the bulb. DISCLOSURE OF THE INVENTION Thus, an object of the present invention is to provide a power supply circuit including a magnetron adapted to supply microwave energy to a microwave discharge light source apparatus including an electrodeless discharge bulb, wherein the circuit is small in size and light in weight; more particularly, an object of the present invention is to reduce the size and weight of the step-up transformer comprised in the circuit. Another object of the present invention is to provide such power supply circuit including a magnetron which supplies microwave energy that is capable of sustaining stable discharge in the electrodeless bulb of the light source apparatus; namely, it is an object of the present invention to provide a power supply circuit which does not cause flickering in the discharge in the bulb and which is capable of sustaining the discharge in the bulb without any fear of extinguishment. According to the present invention, there is provided a microwave discharge light source apparatus with an electrodeless discharge bulb and a circuit system adapted to supply microwave energy to said bulb, said circuit system comprising: first rectifier means, adapted to be coupled to an AC voltage source of a relatively low voltage and frequency, for outputting a rectified voltage of a relatively low voltage; filter means, coupled to said first rectifier means, for smoothing said rectified voltage outputted from said first rectifier means, and for outputting a smoothed rectified voltage; inverter means, coupled to said filter means, for converting said smoothed rectified voltage outputted from said filter means to an AC voltage of a relatively high frequency having a waveform of alternating pulses; a step-up transformer having a primary winding coupled to an output of said inverter means, a secondary winding of the step-up transformer outputting an AC voltage of said relative high frequency and of a relatively high voltage; second rectifier means, coupled to said second winding of said step-up transformer, for rectifying said AC voltage of the relative high frequency and the relative high voltage outputted from said secondary winding to a rectified voltage of a relatively high voltage; and a magnetron coupled to said second rectifier means, to be supplied with and operated by said rectified voltage of the relative high voltage outputted from said second rectifier means; characterised by: pulse width modulation control means for modulating a pulse width of said pulses of said AC voltage outputted from said inverter means; wherein the relative high frequency f, expressed in kiloherz, of the AC voltage outputted by said inverter means is not less than 1500 divided by a diameter D, expressed in millimeters, of said electrodeless discharge bulb: f ≧ 1500 / D. BRIEF DESCRIPTION OF THE DRAWINGS Further details of the invention will become more clear in the following description of the best modes for carrying out the present invention, taken in conjunction with the accompanying drawings, in which: Figures 1a and 1b are schematic sectional views of conventional microwave discharge light source apparatuses; Figures 2a and 2b are diagrams showing conventional power supply circuits for a magnetron, which may be installed to supply microwave energy to an apparatus shown in Figures 1a or 1b; Figure 3a is a diagram showing a power supply circuit according to a first embodiment of the invention which is the subject of European patent application 88906879.7 (EP 0326619) from which this application was divided; Figure 3b is a block diagram showing details of the PWM control circuit in the power supply circuit of Figure 3a; Figure 4 shows waveform of voltages and currents in the circuit of Figure 3a; Fig. 5 shows the curent-voltage characteristic of a magnetron; Fig. 6 shows the relationships between the pulse width of magnitude corresponding to the output power of the mangetron; Fig. 7 shows the relationships between the pulse width of the gate signals supplied to the inverter switching circuit and a magnitude corresponding to the peak magnetron current; Figs. 8 and 9 are diagrams showing power supply circuits for a magnetron according to the second and the third embodiment, respectively, of that invention; Fig. 10 is a diagram showing a power supply circuit for a magnetron according to an embodiment of the present invention; and Fig. 11 shows a waveforms of the magnetron output power in the circuit of Fig. 10. Fundamental Structure and Operation Referring now to Figs. 3a and 3b of the drawings, a first embodiment according to the invention of EP 0326619 is described. The power supply circuit for the magnetron 1 comprises a diode bridge full-wave rectifier circuit 2, the input terminals of which are coupled across a commercially available AC voltage source E, typically on the order of 100 to 220 volts RMS at 50 to 60 Hz. A voltage divider consisting of a pair of resistors R1 and R2 connected in series is coupled across the output terminals of the rectifier circuit 2. Further, a capacitor C1 constituting a smoothing filter circuit is coupled across the output terminals of the rectifier circuit 2 to supply a substantially constant DC voltage therefrom. The input terminals of the inverter switching circuit comprising four MOSFETs (metal oxide semiconductor field effect transistors) Q1 through Q4 connected in bridge circuit relationship are coupled across the output terminals of the filter circuit, the capacitor C1; the output terminals of the switching circuit is coupled across the primary or input winding P of the step-up transformer T having a step-up ratio of 1 to n, a reactor L being inserted in series with the primary winding P. The inverter switching circuit further comprises four diodes D1 through D4 for reverse currents, which are coupled across the source and the drain terminal of the MOSFETs Q1 through Q4, respectively, the gate terminals of the MOSFETs being coupled to the output terminals of the PWM (pulse width modulation) control circuit 3. Further, a voltage doubler half-wave rectifier circuit consisting of a capacitor C2 and a diode D5 connected in series is coupled across the secondary or output winding S of the transformer T; the output terminals of the rectifier circuit, i.e., the terminals across the diode D5, are coupled across the cathode K and the anode An of the magnetron 1 to supply a pulsating DC current I Mg thereto. The output terminals of a current detector 4 for detecting the current flowing through the secondary winding S of the transformer T are coupled to the PWM control circuit 3 to output a voltage Vf corresponding to the current flowing through the secondary winding S. As, shown in Fig. 3b, the control circuit 3 comprises a half-wave rectifier 3a rectifying the output Vf of the current detector 4, a smoothing filter 3b coupled to the output of the rectifier 3a to output a smoothed voltage Vf corresponding to the mean value of the voltage Vf; the error detector or subtractor 3d is coupled to the outputs of the filter 3b and a variable resistor 3c outputting a pre-set reference voltage Vr, and outputs the difference: Ve = Vr - Vf′ between the reference Vr and the mean voltage Vr′. The amplifier 3e amplifies the error or the difference Ve by a factor A, and outputs an amplified error signal: Ve′ = A · Ve. Further, for the purpose of feeding the value of the voltage Vo forward to the control circuit 3, the output terminal of the voltage devider consisting of the resistors R1 and R2 i.e., the terminal at the intermediate position between the two resistors R1 and R2, which outputs a voltage Vin corresponding to the output voltage Vo of the smoothing filter capacitor C1, is coupled to another amplifier 3g which amplifies the signal Vin by a factor of B to output a signal: Vb = B · Vin The subtractor 3f coupled to the outputs of the amplifiers 3e and 3g outputs the difference Vp = Ve′ - Vb to the modulator 3h. The modulator 3h outputs pulses Vw at a predetermined fixed frequency which is substantially higher than that of the AC voltage source E, the width of the pulses Vw being modulated, i.e., varied with respect to a predetermined fixed pulse width, in proportion to the value of the signal Vp. The driver circuit 3i coupled to the output of the modulator 3h outputs gate signals to the MOSFETs Q1 through Q4 of the inverter switching circuit in response to the signal Vw, and alternately turns on and off the MOSFETs Q1 and Q4 and the MOSFETs Q2 and Q3. Thus, high frequency AC current flows through the primary winding P of the transformer T to induce an AC voltage in the secondary winding S thereof, which is rectified and supplied to the magnetron 1 through the rectifier circuit consisting of the capacitor C2 and the diode D5. More explicit description of the operation of the circuit of Figs. 3a and 3b is as follows. First, the operation during a positive half-cycle Tp of the inverter switching cycle is described, referring to Fig. 4 as well as Figs. 3a and 3b. When the driver 3i of the control circuit 3 turns on the MOSFETs Q1 and Q4, while the MOSFETs Q3 and Q4 are turned off, the output voltage V1 of the inverter switching circuit rises substantially to a level equal to the output voltage Vo of the filtering capacitor C1 and is kept thereat during the time interval in which the MOSFETs Q1 and Q4 are tuned on; thus, the output voltage V1 of the inverter switching circuit has a square-shaped waveform, as shown in Fig. 4(a). The duration T ON of the positive voltage V1 i.e., the pulse width thereof corresponds to the pulse width of the gate signal outputted from the driver 3i and that of the signal Vw outputted from the PWM modulator 3h of the control circuit 3; the height of the pulse V1 is substantially equal to the output voltage Vo of the filtering capacitor C1. Due to the inductance of the reactor L connected in series with the primary winding P of the transformer T, the current i₁ flowing through the primary winding P in the direction shown by the arrow in Fig. 3a increases gradually from zero to a maximum during the time in which the voltage V1 is maintained at the positive level, as shown in Fig. 4(b); after the MOSFETs Q1 and Q4 are turned off and the voltage V1 returns to zero level, the current i₁ in the primary winding P of the transformer persists during a short time Tx, due to the existance of the inductance of the reactor L connected in series with the primary winding P. During this short time period Tx, the current i₁ flows through the diodes D2 and D3 to charge the capacitor C1. The current induced in the secondary winding S of the transformer during this positive half-cycle Tp of the inverter has a polarity corresponding to the conducting direction of the diode D5; thus, no currents i Mg flows through the magnetron 1 and the voltage V2 across the cathode K and the anode An of the magnetron 1 is equal to zero, as shown in Fig. 4 (c) and (d), the capacitor C2 being charged by the current induced in the secondary winding S during the positive half-cycle Tp. The operation of the power supply circuit during the negative half-cycle Tn of the inverter is as follows. During the negative half-cycle Tn, the MOSFETs Q2 and Q3 are turned on by the control circuit 3; thus, the polarities of the output voltage V1 of the inverter switching circuit and the current i₁ flowing through the primary winding P of the transformer T are reversed, as shown in Fig. 4 (a) and (b). Except for this, the operation of the circuit electrically coupled to the primary winding P of the transformer T during the negative half cycle Tn is similar to the operation thereof in the positive half-cycle Tp. However, the voltage induced in the secondary winding S by the current i₁ flowing through the primary winding P in the direction opposite to that shown by the arrow in Fig. 3a, the induced voltage in the secondary winding S is superposed on the voltage developed across the capacitor C2 which is already charged in the preceding positive half-cycle Tp; thus, as shown in Fig. 4(c), the voltage V2 applied across the magnetron 1 jumps to the voltage level to which the capacitor C2 has been charged in the previous half-cycle Tp, when the MOSFETs Q2 and Q3 are turned on and the output voltage V1 goes down from zero to a negative level as shown in Fig. 4(a). After this, the voltage V2 applied across the mangetron 1 increases gradually during the time T ON in which the MOSFETs Q2 and Q3 are turned on and the output voltage V1 of the switching circuit is kept at the negative level, due to the gradual decrease of the voltage developed across the reactor L during the same time period T ON . The current i Mg flowing through the magnetron 1, on the other hand, increases gradually from Zero to a maximum, as shown in Fig. 4(d) during the time T ON , due to the current-voltage characteristic of the magnetron 1. Namely, as shown in Fig. 5, the voltage V2 across the magnetron 1 plotted along the ordinate is at a finite voltage level Vz when the magnetron current i Mg plotted along the abscissa begins to flow through the magnetron 1. The magnetron voltage V2 increases linearly from this cut-off voltage Vz to a maximum Vz + ΔVz, as the magnetron current i Mg increases from zero to i R , exhibiting the equivalent series resistance r Mg = ΔVz / i R in the linear relationship range. After the MOSFETs Q2 and Q3 are turned off and the output voltage V1 of the inverter switching circuit returns to zero level, the current i₁ in the primary winding P of the transformer T persists in the short length of time Tx due to the reactor L, during which the magnetron voltage V2 and the magnetron current i Mg decreases and returns to the zero level at the end thereof, as shown in Fig. 4 (c) and (d). The output power of the magnetron 1 is held at a constant level by the modulation of the pulse width T ON of the gate signals applied to the MOSFETs Q1 through Q4 from the control circuit 3. Detailed explanation thereof is as follows. The output power P OUT of the magnetron 1 is approximately given by the product of the mean value of the magnetron current i Mg shown in Fig. 4(d) and the magnetron voltage V2, because the rise ΔVz in the voltage V2 is small compared to the magnitude of the cut-off voltage Vz, as shown in Fig. 5, when the magnetron 1 is operated within the rated current and voltage range. Thus, P OUT is approximated as follows: P OUT ≃ f· V z / n (α² + ω²)L · (2V o - V z /n) · l+a l-a·b (l+b), wherein, the meanings of the symbols are as follows: f: the switching frequency of the inverter, or the frequency of the pulses of the voltage V2 and the current i Mg ; α : (r Mg / n² + Ro) / 2L; ω : (l /LC) - α ; α o : Ro / 2L; ω o : (l / LC) - α o ² Ro: the interior resistance of the voltage source; n: step-up ratio of the transformer T; L: inductance of the reactor L; C: the conversion value of the capacitance of the capacitor C4 in a equivalent circuit in which the capacitor C4 is forming part of the circuit electrically coupled to the primary winding P; T ON : the length of time during which the MOSFETs Q1 through Q4 are turned on, which is equal to the pulse width of the output signals of the control circuit 3, or the pulse width of the voltage V1, as shown in Fig. 4(a); the values of a and b in the equation (1) being given as follows: a = ℓ -α o T ON · l ω o · (-α o sin ω o T ON - ω o cos ω o T ON ); b = ℓ -α o T ON · l ω · (- α sin ωT ON - ω cos ω T ON ). Thus, Fig. 6 shows the relationship between the value Y = l + a l - a·b (l + b) appearing in the right hand side of equation (1) and T ON , in the case where n = 10, C = 0.47 x 10⁻⁸ F, Ro = 2Ω, r Mg = 300Ω. As seen from the figure, the value Y increases as the pulse width T ON increases; provided that the frequency f of the inverter is about 100 kHz and the operating range of the pulse width T ON is approximately from 4 to 5 microseconds, the value Y is approximately in linear relationship with the pulse width T ON . Thus, under these conditions, the increase in the output power P OUT given by equation (1) above is approximately proportional to the increase in the pulse width T ON . On the other hand, the mean voltage signal Vf′, which is obtained from the voltage Vf corresponding to the magnetron current i Mg by rectifying and smoothing it by the rectifier 3a and the smoothing filter 3b as shown in Fig. 3b, is proportional to the magnetron output power P OUT . Thus, when the magnetron output power P OUT decreases, the error signal Ve, the increase of which corresponds to the decrease in the magnetron output power P OUT , increases, because the decrease in the output power P OUT increases, the mean voltage signal Vf′ increases, thereby decreasing the error signal Ve. Thus, the pulse with T ON also decreases to decrease the output power P OUT . Therefore, the magnetron output power P OUT is maintained at a constant level determined by the setting of the variable resistor 3c. Further, the peak or maximum value i Mg max during the stable operation of the magnetron 1 is given, when ωT ON > Z, by: I Mg max = l + a l - ab · ℓ -αz/ω · sin z · (2V o - V Z /n) nωL and, when T ON ≦ Z, by: I Mg max = l + a l - ab · ℓ -αT ON · sin ωT ON · (2V o -V Z /N) nωL , wherein Z = tan-l(ω/α). Fig. 7 shows the relationship Between the value X = I Mg max / (2V o - V Z /n) nωL corresponding to the variable factors in the expression (2) and (2)′ and the pulse width T ON , in the case where n = 10, C = 0.47 x 10⁻⁸ F, Ro = 2Ω, r Mg = 300 Ω. As seen from the figure, the value X is proportional to the pulse width T ON when the inductance L of the reactor L is large enough; for example, in the case where the frequency f of the inverter is around 100 kHz and the pulse width T ON is limited within the range from about 4 to 5 microseconds, the magnetron peak current i Mg max can be represented by a linear equation if the value of L is selected at 8 miceohenries at which the value of X is approximately proportional to the pulse width T ON ; namely, i Mg max is approximated by: i Mg max ≃ K·(2Vo - V2/n)·T ON , wherein K is the proportionality constant determined by the relationship between X and T ON . The output voltage Vo of the filtering capacitor C1 appearing in the right hand side of expression (3) above is subject to variation due to the variation in the AC voltage source E: Vo = V DC + ΔV, wherein V DC represents the pure DC, i.e., constant, component of the voltage Vo and ΔV represents the AC component, i.e., variation, of the voltage Vo. In order to maintain the peak current iMg max given by the approximate equation (3) at a constant level irrespective of the variation ΔV in the voltage Vo, T ON should be varied to satisfy the following equation: T ON = K1 / (2V o - V Z /n) wherein K1 represents an arbitrary proportionality constant. By substituting the right hand side of equation (4) into the right hand side of equation (5) and expanding the right hand side of the equation (5) into Taylor series, i.e., into an infinite sum of the powers of ΔV, wherein the infinitesimal terms of degrees equal to or greater than 2 are neglected, the pulse width T ON is approximately expressed as follows: T ON ≃ K2 - K3 · ΔV, wherein K2 and K3 are constants determined by the values of K1, Vo, V DC , and n. On the other hand, the modulating signal Vp outputted from the subtractor 3f to the PWM modulator 3h is given by: Vp = Ve′ - Vin B, wherein Ve′ is constant in a stable operation and Vin is proportional to the voltage Vo = V DC + ΔV. Thus, the pulse width TON of the signal Vw outputted from the modulator 3h, or that of the gate signals outputted from the driver 3i, can be expressed as follows: T ON = K4 - K5 · ΔV, wherein K4 is a constant determined by the magnitude of the amplified error signal Ve′ and the constant voltage component V DC of the voltage Vo, and K5 is a constant determined by the voltage signal Vin and the amplifying factor B of the amplifier 3g. Therefore, by selecting the values of the constants K4 and K5 in equation (7) in such a way that they agree with the values of the constants K2 and K3 in equation (6), respectively, the peak current i Mg max of the magnetron 1 can be maintained at a constant level irrespective of the variation ΔV in the smoothed DC voltage Vo outputted from the filtering capacitor C1. In this manner, the magnetron peak current i Mg max is held substantially constant even when the AC line voltage Source E fluctuates. In other words, the inverter current flowing through the MOSFETs Q1 through Q4 is stabilized, thereby eliminating the danger of failures thereof. Second and Third Mode: Simplified Inverter Switching Circuits Referring now to Figs. 8 and 9 of the drawings, a second and a third embodiment according to the invention of EP 0326619 having a push-pull type inverter switching circuit are described. Figs. 8 and 9 show a second and a third embodiment of the invention of EP 0326619, respectively, both of which have a structure and operation similar to that of the first embodiment of that invention, except for the inverter switching circuit and the position of the reactor. Thus, a full-wave diode bridge rectifier circuit 2 is coupled across the commercial AC voltage source E, the output terminals of the rectifier circuit 2 being coupled across the series connected resistors R1 and R2 constituting a voltage devider and across the capacitor C1 constituting a smoothing filter. The inverter switching circuit, however, consists of a pair of MOSFETs Q1 and Q2, and diodes D1 and D2 coupled across the source and the drain terminal thereof for reverse currents. In the case of the second embodiment shown in Fig. 8, the source and the drain terminal of the MOSFETs Q1 and Q2 are coupled across the negative terminal of the capacitor C1 and the terminals of the primary winding P of the step-up transformer T, respectively, the positive output terminal of the capacitor C1 being coupled to the center tap 0 of the primary winding P of the transformer T. Thus, in this second embodiment, the reactor L having a function corresponding to that of the reactor L of the first embodiment is inserted in series with the secondary winding S of the transformer T, the capacitor C2 and the diode D3 being coupled in series with the secondary winding S and the reactor L to form a rectifier circuit corresponding to the rectifier current consisting of the capacitor C2 and the diode D5, as in the case of the first embodiment. In the case of the third embodiment shown in Fig. 9, the primary winding of the transformer T is devided into two portions P1 and P2; a mutual inductance M having a pair of magnetically coupled coils M1 and M2 is coupled across the terminals O1 and O2 without dot marks in the figure, the mutual inductance M effecting a function corresponding to that of the reactor L of the first embodiment. Thus, the MOSFETs Q1 and Q2 are coupled across the negative terminal of the capacitor C1 and the dotted terminals O3 and O4 of the windings P1 and P2, respectively; the positive terminal of the capacitor C1 is coupled to the terminal between the two coils M1 and M2 of the mutual inductance M. The circuit coupled to the secondary winding S of this third embodiment is similar to that of the first embodiment. In both second and third embodiment, the voltage devider consisting of the series connected resistors R1 and R2 outputs a voltage Vin corresponding to the output voltage Vo of the capacitor C1 to the PWM control circuit 3; the current detector 4 detects the current flowing through the secondary winding S of the transformer T and output a voltage Vf corresponding thereto to the control circuit 3. The control circuit 3, which has a structure and an operation similar to those of the control circuit 3 of the first embodiment, outputs gate signals alternately to the MOSFETs Q1 and Q2, and alternately turns them on and off, modulating the pulse width thereof. Thus, in the positive half-cycle in which the MOSFET Q1 is turned on and the MOSFET Q2 is turned off, the induced voltage in the secondary winding S of the transformer T has a polarity agreeing with that of the diode D3; consequently, the induced current in the secondary winding S charges the capacitor C2 during the positive half-cycle. In the negative half-cycle, the MOSFET Q2 is turned on, while the MOSFET Q1 is turned off; thus, the polarity of the induced voltage in the secondary winding S is reversed, and is applied across the magnetron 1 together with the voltage developed across the capacitor C2. The resulting voltage V2 causing the current i Mg to flow from the anode An to the cathode K of the Magnetron 1. Fourth Mode: Preferred Inverter Frequency Referring now to Fig. 10 of the drawings, an embodiment according to the present invention is described. The power supply circuit shown in Fig. 10 has a structure similar to that of Figure 8. Thus the input terminals of the diode bridge full-wave rectifier circuit 2 are coupled across the output terminals of the commercial AC voltage source E; the output terminals of the rectifier circuit 2 are coupled across the capacitor C1 constituting the smoothing filter circuit. The inverter switching circuit 5 comprises a pair of MOSFETs Q1 and Q2 and diodes D1 and D2 coupled thereacross in reversed polarity. The MOSFETs Q1 and Q2 are coupled across the negative terminal of the capacitor C1 and the terminals O1 and O2 of the primary winding P of the step-up transformer T; the positive terminals of the capacitor C1 is coupled to the center tap 0 of the primary winding P of the transformer T. The voltage doubler half-wave rectifier circuit consisting of a capacitor C2 and a diode D3 connected in series is coupled across the secondary windings S of the transformer T, to supply pulsing DC voltage V2 to the magnetron 1 provided with a cathode K and an anode An. The filament voltage source 1a for the magnetron 1 is explicity shown in Fig. 10. However, this embodiment is simplified compared with the arrangements of Figures 7 and 9 in certain respects. Namely, a reactor L or mutual inductance M is not necessarily provided in the circuit. Further, a current detector for detecting the current flowing through the secondary winding S of the transformer T is not necessarily provided, and the voltage Vo developed across the capacitor C1 is directly supplied to the control circuit 30 and the driver circuit 31. The control circuit 30 and the driver circuit 31 together correspond to the control circuit 3 of the first through the third embodiment of the EP 0326619 invention. The control circuit 30 may primarily be constituted by TL-494, an IC for switching regulator source, produced by TI company, for example, and outputs Vw1 and Vw2 alternately to the driver circuit 31; the pulse width of these pulses Vw1 and Vw2 can be varied in response to the voltage Vo supplied thereto. The driver circuit 31 outputs gate signals alternately to the MOSFETs Q1 and Q2 in response to the pulses Vw1 and Vw2 to turn them alternately on and off. Thus, current alternately flows through the upper and the lower half of the primary winding P from the center tap 0. Consequently, an AC voltage is induced in the secondary winding S of the transformer T1 which is stepped up by a factor equal to the ratio of the number of turns of the secondary winding S to the number of turns of the primary winding P between the center tap 0 can the terminal O1 or O2 of the transformer T. This AC voltage induced in the secondary winding S is converted into a unidirectional pulsing current by the voltage doubler half-wave rectifier circuit consisting of the capacitor C2 and the diode D3, and is applied therefrom across the magnetron 1; thus, the magnetron is driven by a pulsating current. Consequently, the microwave generated by the magnetron 1 pulsates. Fig. 11 shows the change of the output power P OUT of the microwave generated to time plotted along the abscissa. The reason why the output power P OUT of the magnetron 1 takes the waveform as shown in Fig. 11 is as follows. In the half-cycle of the switching circuit 5 in which the MOSFET Q2 is turned on, the induced voltage in the secondary winding S has a polarity which agrees with the forward direction of the diode D3. Thus, in this half-cycle, the capacitor C2 is charged by the induced current flowing through the diode D3 and the secondary winding S; no voltage is applied across the magnetron 1. In the succeeding half-cycle in which the MOSFET Q1 is turned on while the MOSFET Q2 is turned off, a voltage having a reversed polarity with respect to the diode D3 is induced in the secondary winding S of the transformer T. Thus, the diode D3 is turned off, and the sum of the voltages induced in the secondary winding S and developed across the capacitor C2, which is charged in the previous half-cycle, is applied across the magnetron 1. In Fig. 11, t1 corresponds to the time in which the MOSFET Q1 is turned on, to drive the magnetron 1 by the sum of the induced voltage in the secondary winding S and the voltage developed across the capacitor C2; t2 represents the time in which the MOSFET Q1 is turned off. Thus, the waveform of the microwave output power of the magnetron 1 consists of a train of pulses having a pulse width t1 and recuring at the period To = t1 + t2, as shown in Fig. 11. The magnetron 1 is disposed in a microwave discharge light source apparatus, such as those shown in Figs. 1a and 1b, which comprise a spherical electrodeless bulb. Then, the inverter switching frequency f, i.e. the frequency f = 1/To of the pulses of the microwave output power P OUT of the magnetron 1 expressed in kHz, is preferred to be not less than the magnitude 1500/D; namely; it is preferred that f ≧ 1500/D, wherein D = (the diameter of the electrodeless bulb expressed in millimeters). The reason therefor is as follows. An experiment has been conducted utilizing a microwave discharge light source apparatus shown in Fig. 1a, wherein the bulb 4 has a diameter of 30 mm, 100 mg of mercury being encapsulated therein as an light emitting substance. When the magnetron input power is set at 1.5 kW and the inverter switching frequency f is varied in the range of from about 10 to 20 kHz, the discharge in the bulb become unstable in intervals of substantial widths within this frequency range. This unstability in the discharge is inferred to be due to an acoustic resonance phenomenon similar to that caused by sound waves in the bulb having electrodes, which is clarified in Shomeigakkaishi (Illumination Society Review) vol. 67 No. 2, pp. 55 through 61. However, in the case of a discharge bulb having electrodes, the discharge therein is an arc discharge caused across the two electrodes, the discharging region generally forming a line across the electrodes. In contrast thereto, the bulb which is utilized in the light source apparatus according to the present invention is electrodeless; the discharge therein is maintained by the microwave energy entering thereinto through the wall thereof: when the bulb has a spherical shape as in the apparatus of Fig. 1a, the discharge therein is also spherical. Thus, the state of the discharge caused in the electrodeless bulb by a microwave according to the present invention is completely different from that of the discharge bulb having electrodes; consequently, the acoustic resonance phenomenon of the electrodeless bulb must also differ from that of the bulb having electrodes. More explicitly, it is known that the acoustic resonance phenomenon depends on the velocity of the sound wave in the discharge medium gas and on the dimension and the shape of the discharge bulb; the velocity of the sound wave varies with the temperature and the pressure of the gas through which it is propagated. Thus, as described above, due to the difference in the states of the discharge in the electrodeless bulb and the bulb with electrodes, the temperatures and the temperature distributions of the gas, or the distributions of the velocity of the sound waves in these two types of bulbs, are different from each other. In spite of these differences, certain conclusions may be drawn from the experiments conducted by the inventors. Namely, in an experiment utilizing the apparatus of Fig. 1a having a spherical electrodeless bulb 30 mm across (D = 30 mm), wherein the inverter switching frequency f was varied to test the stability of the discharge in the bulb in varying frequency, it has been observed as follows: when the frequency f is less than or equal to 50 kHz, the intervals of frequency f in which the discharge is unstable occupy considerable proportions; when the frequency f is greater than 50 kHz, however, the widths of these intervals shrinks rapidly as the frequency f is increased. Thus, under the above condition, it can be concluded that the stable discharge can be maintained in the electrodeless bulb if the discharge in the bulb is caused by the microwave generated by a magnetron driven at a switching frequency not less than 50 kHz. From this particular example, general formula for the preferred value of the inverter switching frequency f can be obtained. Namely, the frequency f at which an acoustic resonance phenomenon takes place is proportional to the sound wave velocity C in the discharging gas and inversely proportional to the diameter D of the discharge bulb: f ∝ C / D. The sound wave velocity C in the gas, however, varies little where the mercury in the electrodeless bulb attains a relatively high pressure, i.e. 1 atmosphere, in operation. Thus, the resonating frequency is inversely proportional to the diameter D of the bulb. In the above experiment, it has been decided that the resonance is substantially reduced when the frequency f is not less than 50 kHz at D = 30 mm. Thus, it can be generally concluded that the acoustic resonance causing unstability in the discharge can be substantially reduced if the frequency f satisfies the following inequality: f (kHz) ≧ 1500 / D, wherein D represents the inner diameter of the bulb in millimeters. Further, if the frequency f satisfies equality (8) above, there is no danger that the discharge in the bulb is extinguished in the time intervals t2 between the pulses of the microwave output power shown in Fig. 11, as explained in what follows: In the power supply circuit of Fig. 10, a half-wave voltage doubler rectifier circuit consisting of a capacitor C2 and a diode D3 is used to rectify the voltage induced in the secondary winding S of the transformer T. Thus, as shown in Fig. 11, the microwave output power P OUT is reduced to zero in the time intervals t2 between the time intervals t1 in which the MOSFET Q1 is turned on. The duration of the time intervals t2, however, does not exceed 1 millisecond, provided that the frequency f is not less than 1 kHz, even if the pulse width t1 is decreased in PWM control thereof. On the other hand, the so-called after-glow of the discharge, during which the discharge is maintained after the energy supply thereto ceases, is not less than about 1 milliseconds, provided that the plasma generating medium in the bulb consists of substances usually utilized in a discharge bulb, i.e., a rare gas, or a combination of rare gas and mercury or other metal. Thus, if the length of the time intervals t2 in which no microwave energy is supplied to the bulb does not exceed 1 millisecond, the discharge in the bulb is maintaining through the time interval t2 because, after the supply of the microwave energy carried by a pulse thereof ceases, the discharge in the bulb is maintained by the after-glow until the succeeding pulse of microwave energy is supplied thereto. By the way, if the frequency f satisfies inequality (8) above, the diameter D of the bulb must be as great as 1500 mm to reduce the frequency f to 1 kHz at which the length of the time intervals t2 can not exceed 1 milliseconds. However, the diameter D of the bulb does not exceed 100 mm in practical electrodeless discharge light source apparatus. Thus, if the frequency f satisfies inequality (8), the length of time intervals t2 during which the microwave energy supply ceases does not exceed 1 millisecond in a practical electrodeless discharge bulb; consequently, there is no danger that the discharge is extinguished between the microwave energy supply pulses. In an alternative arrangement of inverter means, four transistors may be electrically connected in a full bridge circuit relationship. In alternative embodiments, the inductance means may be arranged as it is in Figure 9. Additionally, a current detector arranged similarly to the one in Figure 3a may be provided. While description was made of particular embodiments according to the present invention, it will be understood that many modifications may be made without departing from the scope of the appended claims. For example, the inverter switching may be constituted by a half bridge circuit or monolithic forward circuit instead of full bridge circuit or push-pull circuit. Further, the switching circuit may comprise, instead of the MOSFETs utilized in the embodiments described above, power transistors SIT or GTO, SI thyristors, or magnetic amplifiers. Further still, the inductance may be constituted by a leakage inductance of the step-up transformer, ie, the self-inductances of the primary and secondary winding thereof.

Claims: A microwave discharge light source apparatus with an electrodeless discharge bulb and a circuit system adapted to supply microwave energy to said bulb, said circuit system comprising: first rectifier means (2), adapted to be coupled to an AC voltage source (E) of a relatively low voltage and frequency, for outputting a rectified voltage (Vo) of a relatively low voltage; filter means (C1), coupled to said first rectifier means, for smoothing said rectified voltage outputted from said first rectifier means, and for outputting a smoothed rectified voltage; inverter means (5), coupled to said filter means (C1), for converting said smoothed rectified voltage outputted from said filter means to an AC voltage of a relatively high frequency having a waveform of alternating pulses; a step-up transformer (T) having a primary winding (P) coupled to an output of said inverter means (5), a secondary winding (S) of the step-up transformer outputting an AC voltage of said relative high frequency and of a relatively high voltage; second rectifier means (C2, D3), coupled to said second winding (S) of said step-up transformer, for rectifying said AC voltage of the relative high frequency and the relative high voltage outputted from said secondary winding to a rectified voltage of a relatively high voltage; and a magnetron (1) coupled to said second rectifier means (C2, D3), to be supplied with and operated by said rectified voltage of the relative high voltage outputted from said second rectifier means; characterised by: pulse width modulation control means (31) for modulating a pulse width of said pulses of said AC voltage outputted from said inverter means; characterised in that the relative high frequency f, expressed in kiloherz, of the AC voltage outputted by said inverter means (5) is not less than 1500 divided by a diameter D, expressed in millimeters, of said electrodeless discharge bulb: f ≧ 1500 / D. Apparatus as claimed in Claim 1, further comprising inductance means, operatively coupled to said step-up transformer, for suppressing a rapid change in a level of a current flowing through a winding of said step-up transformer (T). Apparatus as claimed in Claim 1 or Claim 2, wherein said inverter means (5) comprises a switching circuit including four transistors electrically connected in full bridge circuit relationship. Apparatus as claimed in Claim 1 or Claim 2, wherein said inverter means (5) comprises a switching circuit including a pair of transistors (Q1, Q2) electrically connected in push-pull circuit relationship. Apparatus as claimed in any one of Claims 2, 3 or 4, wherein said inductance means comprises an inductance electrically connected in series with said primary winding of said step-up transformer. Apparatus as claimed in any one of Claims 2, 3 or 4, wherein said inductance means comprises an inductance electrically connected in series with said secondary winding of said step-up transformer. Apparatus as claimed in any one of the Claims 2 through 6, wherein said inductance means comprises a leakage inductance of said step-up transformer. Apparatus as claimed in claim 2 or 6, wherein said primary winding of said step-up transformer (T) comprises a first and a second winding portion, and said inductance means comprises a mutual inductance electrically connected between said first and second winding portion of said primary winding in series circuit relationship. Apparatus as claimed in any one of Claims 1 to 8 wherein said pulse width modulation control means (31) comprises current detector means for detecting a current level of a current flowing through said magnetron, and means for varying said pulse width of said AC current outputted by said inverter means in response to said current level of the current flowing through the magnetron detected by said detector means, thereby maintaining an output power of the magnetron at a predetermined level. Apparatus as claimed in claim 9 , wherein said predetermined level is variable. Apparatus as claimed in any one of Claims 1 to 10, wherein said first rectifier means (2) comprises four diodes electrically connected in bridge circuit relationship. Apparatus as claimed in any one of claim 1 to 11, wherein said filter means comprises a capacitor (C1) electrically connected across output terminals of said first rectifier means. Apparatus as claimed in any one of Claims 1 to 12, wherein said second rectifier means comprises a diode (D3) and a capacitor (C2) electrically connected in a series coupled across terminals of said second winding (S) of the step-up transformer.

Inventors: IWATA AKIHIKO C O MITSUBISHI D, KODAMA HITOSHI C O MITSUBISHI, MAGOME KAZUO C O MITSUBISHI DE, SHODA ISAO C O MITSUBISHI DENK, TAKI MASAKAZU C O MITSUBISHI D, YOSHIZAWA KENJI C O MITSUBISHI, IWATA, AKIHIKO, C/O MITSUBISHI DENKI K. K., KODAMA, HITOSHI, C/O MITSUBISHI DENKI K. K., MAGOME, KAZUO, C/O MITSUBISHI DENKI K. K., SHODA, ISAO, C/O MITSUBISHI DENKI K. K., TAKI, MASAKAZU, C/O MITSUBISHI DENKI K. K., YOSHIZAWA, KENJI, C/O MITSUBISHI DENKI K. K.
Applicants: MITSUBISHI ELECTRIC CORP, MITSUBISHI DENKI KABUSHIKI KAISHA
Application year: 1988
Main classification: H05B41
IPC codes: H01J65, H05B41