TTL-TO-ECL INPUT TRANSLATOR/DRIVER CIRCUIT

Abstract: The present invention provides a high speed low power electrical circuit for converting true TTL level signals to true ECL level signals. The circuit only has a single buffer delay with some small addi­ tional delay due to an input emitter follower stage. The circuit includes a clamped, switched emitter fol­ lower (R1, Q1, Q2, Q3) which acts as a level shifting comparator; a selfcentering reference threshold translator (Q7, Q8, Q10, Q15, R5, R6, R12); a clamped level shifted input translator (Q4, Q5, Q14, R3, R4, R10); and, an ECL Buffer Driver (Q11, Q12, Q13, R7, R8, R11). The circuit also includes a TTL reference (D2, Q6, Q9, R2) and an ECL reference (Q7, Q8, R5, R6) which are tied together. If the TTL reference level shifts slightly due to temperature changes, supply voltage shifts or other factors. the ECL voltage refer­ ence will automatically shift by an appropriate per­ centage to compensate for the original shift in the TTL reference.

Description: The present invention relates to electronic circuitry and more particularly to voltage level translator circuitry. Two well known and commonly used types of electronic logic circuitry are TTL circuitry and ECL circuitry. TTL circuitry is designed to switch between a low voltage level that ranges from 0.0 volts to 0.8 volts and a high voltage that ranges from 2.0 volts to 5.5 volts. ECL circuitry operates at much lower voltage levels. The type of ECL circuitry of interest to this invention is conventionally designed to switch between voltage levels of minus 1.5 volts and minus 0.8 volts, that is, between -1500 millivolts and -800 millivolts. Many circuits can be designed to translate TTL signals to ECL signals. However, the known circuits for translating signals from TTL to ECL levels are relatively slow and they operate at relatively high power levels. One well known and commonly used translator circuit, for instance as known from document WO-A-86/02792, compares the TTL input signals to a reference voltage. A pseudo-ECL current switch provides true/false output values at pseudo-ECL voltage levels. These pseudo-ECL voltage levels are fed via emitter followers to a resistor ladder. The resistor ladder is differentially tapped at the middle to drive a true ECL current switch which in turn provides normal collector based true/false outputs which can serve as inputs to emitter followers. Such circuitry in effect has two buffers connected in series which introduce a relatively long delay into the signal propagation path. IEEE Intercon Conference vol.6, 1975, pp 18/3 1-8; J Buie "3D-LSI Compatible Logic Family" discloses a TTL to EFL translation in which the output is taken from a circuit node at the emitters of two TTL input transistors. The output is taken through a level shifting buffer to an EFL output terminal. The object of the present invention is to provide an improved TTL level to ECL level translator. Another object of the present invention is to provide a high speed TTL level to ECL level translator. Yet another object of the present invention is to provide a TTL level to ECL level translator which operates at a relatively low power level. Still another object of this invention is to provide a high speed low power TTL level to ECL level translator. A still further object of this invention is to provide a circuit which performs, precise, controlled voltage translations under a wide variation of fabrication, process, temperature, voltage, etc. variations. According to the present invention there is provided a high-speed low power voltage translation circuit for translating input signals that vary between relatively large first and second positive voltages into output signals that vary between relatively small third and fourth negative voltages, comprising a reference voltage generator for providing a first positive voltage threshold reference having a value between said first and second voltages; a clamped level shifting comparator, having first and second emitter-coupled input transistors responsive to said input signals and said first threshold reference respectively for generating an intermediate voltage at a circuit node formed at the common emitter connection of said input transistors, said intermediate voltage being positive and having a relatively small voltage swing; a reference threshold translator, for generating a second voltage threshold reference having a value between said third and fourth negative voltages; a current source; a voltage translator connected in series between the circuit node and the current source; and buffer driver means responsive to the translated voltage from the voltage translator and to said second voltage threshold reference for generating said output signals. The present invention will be further described hereinafter with reference to the following description of an exemplary embodiment and the accompanying drawings, in which: Figure 1 is a circuit diagram of a circuit built in accordance with the present invention. Figure 2A shows the TTL input voltage levels. Figure 2B shows the voltage levels at node 1. Figure 2C shows the voltage levels at node 2. Figure 2D shows the ECL voltage levels. The circuit shown in Figure 1 includes fifteen transistors Q1 to Q15, three diodes D1 to D3, and thirteen resistors R1 to R13. The purpose of the circuit is to convert from true TTL voltage levels to true ECL voltage levels. The voltage levels at the input of the circuit are shown in Figure 2A and the voltage at the output is shown in Figure 2D. The voltage levels at the intermediate nodes 1 and 2 are shown in Figures 2B and 2C. The voltage levels at the output of the circuit are shown in Figure 2D. The cross-hatched lines in Figure 2A represent an allowable voltage range. The Figure is not meant to be to scale. The range is 800 millivolts on the low side and 3 volts on the high side. Figures 2B, 2C and 2D do not show any cross-hatched ranges since at the points illustrated by Figures 2B, 2C and 2D the ranges are in the order only 100 millivolts and this is too small to show in the Figures. The specific value of the resistors is not particularly important. What is significant is the ratio of certain resistors as will be explained in detail later. Diode D2, transistors Q6 and Q9 and resistor R2, provide an Input reference voltage threshold. In Figure 2 this is labeled TTL reference. Resistor R1, and transistors Q1, Q2 and Q3 form a clamped level shifting comparator. These circuits produce a well controlled small voltage swing at node 1 in response to the relatively large input voltage changes. In Figure 2B this voltage is labeled intermediate voltage at node 1. Transistors Q7, Q8, Q10 and Q15 and resistors R5, R6 and R12 provide a self-centering reference threshold translator. This part of the circuit maintains a relationship between the TTL reference voltage shown in Figure 2A and the ECL voltage level shown in Figure 2C. If the TTL voltage reference changes due to changes in temperature or due to other factors, the ECL voltage will change by a fixed percentage of the change in the TTL reference. Thus, the ECL reference at node 3 has the identical dependencies for change in variables such as temperature, supply voltages and process variables as the translated ECL input voltages provided at node 2. Transistors Q4, Q5, Q14 and resistors R3, R4 and R10 provide a resistor ratioed input voltage translation which precisely lowers the voltage which appears at node 1. Transistors Q11, Q12 and Q13 together with resistors R7, R8, and R11 provide an ECL buffer driver which responds to the level shifted signals and which drives the output. The operation of each part of the circuit will now be described in detail. TTL Input Reference Voltage Threshold : Devices D2, Q6, Q9 and R2 define the input reference threshold for the TTL input circuit. This threshold is shown in Figure 2A and labeled TTL reference. Its value is calculated as follows: VCC - Iref * R2 - Vbe(Q9) - Vbe(Q6) - Vsd(D2) = 0 Iref = [VCC - Vbe(Q() - Vbe(Q6) - Vsd(D2)] / R2 Let Vbe(Q6) = Vbe(Q7) = Vbe and Vsd(D2) = Vsd(D1) = Vsd, by design. Vref = Vsd + 2Vbe, and V(Q2) = Vref. In this way, this circuit section provides the base of Q2 with Vref =∼ 2Vbe+Vsd. Note that the true reference voltage as observed at the input is 2Vbe, since Vsd(D2) and Vsd(D1) cancel. The performance of this circuit across the voltage, temperature stress envelope tracks the behavior of the conventional circuit. Thus, this reference introduces no new system level electrical design constraints. TTL Input Clamped Level-Shifting Comparator : Devices R1, Q1, Q2 and Q3 are used to develop a relatively small, well controlled ECL-like voltage at the emitter-tie noted labelled node 1, from the large TTL input voltage transitions. The voltage at node 1 is shown in Figure 2B. The voltage at node 1 is generated as follows. Let the input "A" be at VIL, where, with respect to TTL GND, VILmin = 0.0 V <= VIL <= 0.8 V = VILmax Then, D1 conducts IIL current, which is defined by, IIL = [VCC - Vsd(D1) - VIL] / R1 Consequently, the base of Q1 attains a voltage, Vb(Q1) = VIL + Vsd. (iii) From the analysis presented above (Eq. ii) it should be noted that, Vb(Q2) = Vref = Vsd + 2Vbe. Since 2Vbe is a higher voltage than VILmax, Vb(Q1) is lower than Vb(Q2). Therefore, transistor Q2 develops a full Vbe, and goes into forward-active conduction, while transistor Q1 attains < 0.5Vbe, and is considered cut-off. Hence, V(1) follows Vb(Q2), and the current source Ics1 (Q14/R10) is satisfied by Q2, via R3, R4 and Q5. In this way, the V(1) low level is established. V(1)LOW = Vbe + Vsd. (iv) Next, let the input "A" be at VIH, where, with respect to TTL GND, VIHmin = 2.0 V <= VIL <= 5.0 V = VIHmax (v) Since VIHmin is a higher voltage than 2Vbe, D1 conducts reverse-leakage current only. IIH current is conducted through R1 to Q3 into the input reference voltage circuit. IIH = [VCC -Vbe(Q3) - Vref] / R1 Transistor Q3 acts as a clamp on Vb(Q1), such that with the input at any voltage greater than VIHmin, e.g., 3Vbe, Vb(Q1)max = Vbe(Q3) + Vref = Vbe(Q3) + 1Vb3 + Vsd (vi) Vb(Q1)max = 3Vbe + Vsd (vii) From the analysis presented above (Eq. ii) it should be noted that, Vb(Q2) = Vref = Vsd + 2Vbe. Vb(Q1) is a higher voltage than Vb(Q2). Therefore, transistor Q1 develops a full Vbe, and goes into forward-active conduction, while transistor Q2 attains < 0.5Vbe, and is considered cut-off. Hence, V(1) follows Vb(Q1), and the current source Ics1 (q14/R10) is satisfied by Q1, via R3, R4 and Q5. In this way, the V(1) high level is established. V(1)HIGH = 2Vbe + Vsd. (viii) It should be clear from the operation of the circuit, as explained above, that this circuit is substantially different from the conventional TTL input comparator. The conventional circuit utilizes the same comparator reference, but converts input voltage into pseudo-ECL levels, and then later converts these to true-ECL using the conventional resistor ladder approach. In the novel technique developed in this document, the input stage can be described as a clamped, switched emitter-follower. Self-Centering Reference Threshold Translator: This circuit includes devices Q7, Q8 and reactors R5 and R6. It translates the TTL reference shown in Figure 2A to the ECL reference shown in Figure 2C. The circuit maintains a relationship between the reference signals over changes in temperature, supply voltage, and changes in process variables. That is, whenever the TTL reference changes, the ECL reference changes by a fixed percentage of the change in the TTL reference. The operation of the Vref-translator circuit is as follows. Let ICs3 represent the current required to satisfy the current source setup by Q15 and R12. Ics3 = [VCS - Vbe(Q15)] / R12 Ics3 is sourced by the parallel combination of transistor Q7 and resistor R6. The Ics3 value must be chosen to provide adequate current to bias transistor Q7 in the forward-active mode, while also allowing I(R6) to be satisfied. To first order, I(R6) = Vbe(Q7) / R6 and, Ie(Q7) = Ics3 - I(R6). Note that Vbe(Q7) is in fact a function of Ie(Q7). V(R5) = R5 * I(R6) + [Ie(Q7) / (Hfe + 1)] where Hfe is the current gain of Q7. Therefore, by an appropriate choice of R5, we can set up an appropriate amount of total voltage level shift provided by this circuit. Let us choose, R5 : R6 = 2.5 : 1. (ix) Then, if we define Hfe to be a large positive number, V(R5) = 2.5 * V(R6) = 2.5 * Vbe(Q7) (x) Therefore, the total voltage translation from the emitter of transistor Q10 to the base of transistor Q12 is set to be 3.5Vb3. From our earlier discussion (Eq. ii), Vref = Vsd + 2Vbe. Hence, applying the voltage translation derived above, we find that, Vb(Q12) = Vref - Vbe(Q10) - 3.5Vbe Vb(Q12) = Vsd + 2Vbe - Vbe - 3.5Vbe, or, Vb(Q12) = Vsd - 2.5Vbe (xi) Note the unusual configuration of transistor Q8. The base-emitter and base-collector junctions are reverse biased in parallel. In this way, a large value, space efficient capacitor is obtained. In order to satisfy the current source Ics2, transistor Q12 must build up incremental base charge to attain an appropriate Vbe value. The Q8 capacitor, if sized properly, can transmit an appropriate amount of charge to the base of Q12 much faster than the R6, Q7 combination can, thus providing significant delay performance enhancement. Clamped Level-Shifted Input Translator : Incoming TTL input signals are converted into a Vbe voltage delta at node 1, as described earlier. In this section, the operation of the V(1)-translator circuit is described. Let Ics1 represent the current required to satisfy the current source setup by Q14 and R10. Ics1 = [VCS - Vbe(Q14)] / R10 Ics1 is sourced by the parallel combination of transistor Q5 and resistor R4. The Ics1 value must be chosen to provide adequate current to bias transistor Q5 in the forward-active mode, while also allowing I(R4) to be satisfied. To first order, I(R4) = Vbe(Q5) / R4 and, Ie(Q5) = Ics1 - I(r4). Note that Vbe(Q5) is in fact a function of Ie(Q5). V(R3) = R3 * I(R4) + [Ie(Q5) / (Hfe + 1)] where Hfe is the current gain of Q5. Therefore, by an appropriate choice of R3, we can set up an appropriate amount of total voltage level shift provided by this circuit. Let us choose, R3 : R4 = 3 : 1. (xii) Then, if we define Hfe to be a large positive number, V(R3) = 3 * V(R4) = 3 * Vbe(Q5) (xiii) Therefore, the total voltage translation from node 1 to the base of transistor Q11, is set to be 4Vbe. From our earlier discussion (Eq. ii), Vref = Vsd + 2Vbe. Hence, applying the voltage translation derived above, we find that, Vb(Q11) = V(1) - 4Vbe (xiv) Applying the earlier V(I)HIGH and V(1)LOW limits, (Eq. iv and viii) we find that, Vb(Q11)HIGH = V(1)HIGH - 4Vbe = Vsd - 2Vbe, (xv) and, Vb(Q11)LOW = V(1)LOW - 4Vb3 = Vsd - 3Vbe (xvi) Comparing Eqs. xv and xvi v. Eq. xi, we find that the original incoming TTL input VIL, VIH voltage values have been interpreted into ECL-type voltage values Vb(Q11)HIGH or Vb(Q11)LOW. Further, an appropriate reference, with excellent inherent tracking, across wide voltage, temperature and process variations, has also been created. Transistor Q4 has a rather unusual configuration. The base-emitter and base-collector junctions are reverse biased in parallel. In this way, a large value, space efficient capacitor is obtained. When an input transition is detected, the Q4 capacitor, if sized properly, transmits an appropriate amount of charge to the base of Q11 much faster than the R4, Q5 combination can, thus providing significant delay performance enhancement. The output emitter followers Q17 / Ioef1 and Q16 / Ioef2 perform the normal voltage level-shifting and provide current drive capability. ECL Buffer Driver : The key to the optimal performance of this circuit lies in the creation of the Vb(Q11) and Vb(Q12) voltages as described above. Once those appropriately conditioned voltages are available, the ECL current switch operates in the normal manner. Resistors R7, R8 and R11, and transistors Q11, Q12 and Q13 form the ECL current switch. If Vb(Q11) > Vb(Q12), then Ics2 is satisfied from ECL VCC via R7, and Vb(Q17) attains an ECL LOW output level, while Vb(Q16) attains an ECL HIGH level. Conversely, if Vb(Q11) < Vb(Q12) then Ics2 is satisfied from ECL VCC via R8, and Vb(Q16) attains an ECL LOW output level, while Vb(Q17) attains an ECL HIGH level.

Claims: A high-speed low power voltage translation circuit for translating input signals (A) that vary between relatively large first and second positive voltages into output signals (YA, YA ) that vary between relatively small third and fourth negative voltages, comprising a reference voltage generator (R2, Q9, Q6, D2) for providing a first positive voltage threshold reference (Vref) having a value between said first and second voltages; a clamped level shifting comparator (R1, D1, Q1, Q2, Q3), having first and second emitter-coupled input transistors (Q1, Q2) responsive to said input signals (A) and said first threshold reference (Vref) respectively for generating an intermediate voltage at a circuit node (1) formed at the common emitter connection of said input transistors (Q1, Q2), said intermediate voltage being positive and having a relatively small voltage swing; a reference threshold translator (Q7, Q8, Q10, Q15, R5, R6, R12) for generating a second voltage threshold reference having a value between said third and fourth negative voltages; a current source (Q14, R10); a voltage translator (Q4, Q5, R3, R4) connected in series between the circuit node (1) and the current source (Q14, R10); and buffer driver means (Q11, Q12, Q16, Q17) responsive to the translated voltage from the voltage translator and to said second voltage threshold reference for generating said output signals (YA, YA ). A voltage translation circuit according to claim 1, wherein the clamped level shifting comparator (R1, D1, Q1, Q2, Q3) operates as a TTL input comparator and includes a schottky diode (D1) connected to communicate the input signals (A) to the first and second input transistors (Q1, Q2). A voltage translation circuit according to claim 1, wherein said voltage translator (Q4, Q5, R3, R4) includes first and second resistors (R3, R4) forming a series connection between said circuit node (1) and said current source (Q14, R10). A voltage translation circuit according to claim 3, wherein said voltage translator (24, Q5, R3, R4) includes third and fourth transistors (Q4, Q5), said third transistor (Q5) having its collector connected to said circuit node (1), its emitter connected to said current source (Q14, R10), and its base connected to the mid-point of said third and fourth resistors (R3,R4) said fourth transistor (Q4) having both its emitter and its collector connected to said circuit node (1), and its base connected to said current source (Q14, R10). A voltage translation circuit according to claim 1, wherein the clamped level shifting comparator (R1, D1, Q1, Q2, Q3) includes a diode (Q3) connected to the bases of said first and second transistors (Q1, Q2), the base of said first transistor (Q1) being connected to receive the input signals (A). A voltage translation circuit according to claim 5, wherein the collectors of the first and second transistors (Q1, Q2) are connected together and to a positive supply voltage. A voltage translation circuit according to any one of the preceding claims, wherein said buffer driver means comprises third, fourth and fifth resistors (R7, R8, R11), and fifth, seventh and eighth transistors (Q11, Q12, Q13) connected to form an ECL current switch, a first side (3) of said switch being connected to said ECL reference, and a second side (2) of said switch being connected to said circuit node (1) through said voltage translator (Q4, Q5, R3, R4). A voltage translation circuit according to any one of the preceding claims, wherein a first intermediate voltage is produced at the circuit node (1), and the voltage translator is operable to translate the first intermediate voltage to a second intermediate voltage, and wherein the buffer driver means is responsive to the second intermediate voltage and the second voltage threshold to generate said output signals.

Inventors: KHAN, AURANGZEB K.
Applicants: TANDEM COMPUTERS INCORPORATED
Application year: 1988
Main classification: H03K19
IPC codes: H03K19