OSCILLATOR NETWORK AND QUADRATURE NETWORK FOR RADIO RECEIVER

Abstract: A sinewave oscillator and a quadrature phase shift network fabricated as an integrated circuit comprising, in order that the network and oscillator be simply adjustable to provide an accurate 90° phase shift and a sinewave of accurate fre­ quency and high spectral purity, a first capacitance means (C1, C2; C4, C5) coupled to the input of a gyrator (G1; G2) comprising first and second amplifiers (10, 12; 22, 24) connected in feedback configuration with a capacitance (C3; C6) connected to the output of the first amplifier. The oscillator network may be adjusted by current control of transconduct­ ance of the amplifiers, and the amplifiers include input signal limiting means to limit the input signal to a linear operating region. An output load resistance (R) may be simulated using a further amplifier connected in feedback configuration.

Description: This invention relates to electrical circuits for use in radio receivers fabricated as integrated circuits. There has recently been proposed communication networks known as radio paging networks which will operate over a very large area, eventually world-wide, and which transmit digital radio paging information to individual network subscribers on a time-slot basis on a subcarrier of local VHF stereo FM entertainment broadcast channels. With a pilot carrier of 19KHz and stereo bands extending from 0-53KHz, the subcarrier has a band width of 9.5KHz centred at 66.5KHz. The subcarrier is received by a subscribers receiver, which may be incorporated in a wrist-watch as an integrated circuit. The type of receiver employed is the Barber receiver which has a principal advantage of high image rejection. The Barber receiver configuration is shown in Figure 1 comprises an aerial 2, for receiving an incoming signal having a carrier frequency fc, coupled to an AGC controlled preamplifier 4. The incoming signal is then fed to two arms 6, 8, where it is mixed in mixers 10, 12 with signals, in phase quadrature with one another, from a local oscillator 14. The frequency of the local oscillator is identical with the carrier frequency fc. The mixed signals in the two arms are fed via coupling capacitors to low-pass filters 16, 18, and amplifiers 20, 22 to further mixers 24, 26 where the signals are mixed with signals, in phase quadrature with one another from an intermediate frequency oscillator 28. The signals in arms 6, 8, now raised in frequency by the i f frequency are summated at 30, limited in amplitude by a limiting amplifier 32, and then demodulated by a detector 34 to yield the digital paging signal. The principal advantage of such a receiver is that of image rejection, since the carrier and local oscillator frequencies are identical and result in a spectrum folded about a base DC level. Further advantages are that the receiver is suitable for integration, having a low external component count, and that on-chip filtering is possible since the selectivity is defined at a low frequency. A principal disadvantage is that the receiver is highly sensitive to phase or gain imbalances in the arms 6, 8 and it is therefore important that the signals provided by the local oscillator and intermediate frequency oscillator are accurately in phase quadrature, to within 1°. Further it is important that the local oscillator and the intermediate frequency oscillator produce signals of accurately defined frequency and high spectral purity. Finally it is normally necessary to track the frequency of the intermediate frequency oscillator with the cut-off frequency of the low-pass filters in the arms, and it is necessary to maintain the phase quadrature relationship while the frequency is varied. So far as concerns the problem of providing signals which are accurately in phase quadrature, there are many types of networks which will provide a 90° phase shift, for example R-C phase shift networks or networks employing inductances. However problems arise when such networks are incorporated in an integrated circuit. With R-C networks there are large tolerances in the values of the integrated capacitors and it is therefore very difficult to preset the network and obtain an accurate phase shift. With inductive networks, inductances are difficult to fabricate in an integrated circuit and are difficult to preset in value. US patent documents US-A-4 193 033 and US-A-3 539 943 disclose integratable inductive circuits including gyrators. It is an object of the present invention to provide a quadrature network which provides an accurately defined 90°C phase shift and which can be fabricated as an integrated circuit without difficulty. In a first aspect, the present invention provides a quadrature network fabricated as an integrated circuit and comprising a first capacitive means including first and second capacitors connected in series between an input port of the network and a point of reference potential, and gyrator means having its input connected across said second capacitor and comprising a first amplifier whose output is coupled to a second point of reference potential by way of second capacitive means and to the input of a second amplifier whose output is coupled in a feedback loop to the input of the first amplifier, the transconductances of the first and second amplifier being controllable such that a signal of a predetermined frequency applied to the input port of the quadrature network is shifted in phase by 90° at the input of the first amplifier, which forms the output port of the quadrature network. By providing for control of the transconductance of the amplifiers it is possible to compensate for variations in capacitance values without the necessity for further processing operations, such as capacitor trimming, on the integrated circuit. Thus an accurately defined phase shift of 90° may be obtained at a predetermined frequency. Further in accordance with the invention, it is unnecessary to fabricate circuit elements such as inductances which do not lend themselves simply to integration. The amplifiers employed preferably include emitter coupled bipolar transistors. One problem with such amplifiers is that there is a very limited input signal range, about 60 mV, in which the output signal is linearly related to the input signal and in which the gyrator will function accurately. It is therefore necessary to provide signal limiting means in order to limit the input signals to the amplifier to the linear regions. So far as concerns the problem of providing a sinewave oscillator whose frequency may accurately be determined and which provides a signal of high spectral purity and which may be fabricated as an integrated circuit, there are two common types of sinewave oscillators which may be mentioned, namely phase shift oscillators and oscillators employing an inductive-capacitive reactive network in a feedback loop. The problem with phase shift oscillators in integrated circuits is that the frequency is not accurately controllable since the capacitance values employed vary within wide limits and it is necessary to perform trimming operations to obtain predictable frequency values. The problem with inductive-capacitive networks in sinewave oscillators in integrated circuits is that inductors are difficult to fabricate on chip and even more difficult to adjust in value. In order to meet this problem, the present invention provides, in a second aspect, a sinewave oscillator circuit fabricated as an integrated circuit, the oscillator comprising a main amplifier having its output coupled to its input in a feedback loop which includes a reactive network providing a required phase shift to sustain oscillations, the reactive network comprising a first capacitive means and a gyrator, the gyrator comprising a first amplifier whose output is coupled to a second capacitive means and to the input of a second amplifier whose output is coupled in a feedback loop to the input of the first amplifier, and the transconductances of the first and second amplifiers being controllable to control the impedance of the gyrator and thereby the frequency of oscillation of the oscillator circuit. It is possible to adjust the gyrator impedance simply in order to vary the frequency of the oscillator or to compensate for variations in capacitance value. Further the gyrator impedance may be defined to within close limits in order to provide an oscillator signal of high spectral purity. All of the components of the oscillator circuit may simply be integrated. One problem which arises is that it is necessary for the main amplifier to operate in a non-linear region so that it will saturate in order to limit the amplitude of the oscillations. However, the amplifiers employed in the gyrator must operate in a linear region in order to ensure an accurately predictable gyrator impedance. The type of amplifier preferably employed in the gyrator comprises emitter coupled bipolar transistors which may have a linear operating region for as small as 60mV input voltage. It is therefore necessary to include further amplifying stages and regulate the gain in the various parts of the oscillator circuit to ensure that voltage signals of an apprporiate magnitude are applied to the amplifiers of the circuit. So far as concerns the problem of maintaining an accurately defined quadrature phase shift during variation of oscillator frequency, naturally the problems arise which have been mentioned above with regard to the first and second aspects of the invention. In addition it is necessary to provide a simple and accurate method of adjusting a phase shift network with adjustment of the oscillator frequency. This may be achieved by using the quadrature network according to the first aspect of the invention in combination with the oscillator according to the second aspect of the invention and ganging together the transconductance control inputs of the gyrator amplifiers. Thus the present invention provides, in a third aspect, an electrical circuit comprising an oscillator coupled to a quadrature phase shift network, the oscillator including a main amplifier having a feedback loop including a reactive network, the reactive network including a first capacitive means and a first gyrator, the first gyrator comprising a first amplifier whose output is coupled to a second capacitive means and to the input of a second amplifier whose output is coupled in a feedback loop to the input of the first amplifier, and the quadrature network including a third capacitive means coupled to a second gyrator, the second gyrator comprising a third amplifier whose output is coupled to a fourth capacitive means and to the input of a fourth amplifier whose output is coupled in a feedback loop to the input of the third amplifier, the arrangement being such that a signal applied at a predetermined frequency to the input of the quadrature network is shifted in phase by 90° at the output of the quadrature network, and each of said first, second, third and fourth amplifiers including means for controlling the transconductance of the amplifier, said controlling means being ganged together whereby to vary the frequency of the oscillator while maintaining a phase shift of 90° from the quadrature network at the oscillator frequency. Thus the electrical circuit according to the present invention provides a simple and convenient method of providing signals accurately maintained in phase quadrature, at a precisely defined frequency over a range of frequencies. A preferred embodiment of the invention will now be described with reference to the accompanying drawings, wherein:- Figure 1 is a circuit diagram of a Barber Receiver in which the present invention may be incorporated; Figure 2 is a schematic circuit diagram of the preferred embodiment of an electrical circuit according to the invention; Figure 3 is a more detailed circuit diagram of the internal construction of each of the gyrator amplifiers of Figure 2; Figure 4 is a graph indicating phase shift and signal gain as a function of frequency for the quadrature network; Figure 5 is a circuit diagram of a modification of the phase quadrature section of Figure 2; and Figure 6 is a schematic circuit diagram of the oscillator section of Figure 2, showing gain factors of the amplifying stages. Referring now to the drawings, Figure 2 shows an electrical circuit comprising an oscillator section 2 and a quadrature phase shift network 4. Oscillator section 2 is based on the standard Colpitts configuration and comprises a main amplifier 6 having its output connected to its input in a positive feedback loop 8, the feedback loop including a reactive network comprising series coupled capacitors C1, C2 connected between ground and the input to amplifier 6 and a gyrator G1 connected across capacitors C1, C2. Gyrator G1 comprises amplifiers 10, 12 with amplifier 10 being a non-inverting amplifier whose output is connected to one plate of a capacitor C3 whose other plate is connected to ground and whose output is connected to the input of amplifier 12, which provides an inverted output, the inverted output being fed back to the input of amplifier 10 thereby to provide a gyrator configuration. Quadrature phase shift network 4 comprises input and output ports In and Out . A capacitive network coupled to the input port comprises a series connected capacitance C4 and a capacitance C5 connected to earth. An output signal is developed across a load resistance R connected in parallel with capacitance C5. A gyrator G2 is connected to the output port node and comprises a first non-inverting amplifier 20 having a mutual conductance +g m with its output connected to one plate of a capacitance C6, the other plate of capacitance C6 being connected to ground. The output of amplifier 20 is also connected to the input of a second inverting amplifier 22 having a mutual conductance -g m , the output of amplifier 4 being connected in a feedback loop to the input of amplifier 2. It may be shown that for an input signal of frequency ω the impedance Z of the gyrator means is given by:- Z = jω C6/ g ² m . Further as may be seen from the graph of Figure 4, there is a resonance in the signal gain |V₀|V i | at a frequency ω, where:- ω²=[g m ²/C6 (C4 + C5)]⁻¹ At this resonance frequency there is a phase shift  between the signals at the input and output ports of 90°. Thus it may be seen by controlling the transconductance of the amplifiers by suitable adjustment of a control current, it is possible accurately to preset the parameters of the quadrature network, even where the network is fabricated as an integrated circuit with consequent wide variations in the values of the capacitors. An amplifier circuit shown in more detail in Figure 3, comprises a pair of emitter coupled bipolar transistors 36, 38 coupled by a constant current source 40 and having constant current collector loads 42, 44. The output from transistor 36 provides a positive transconductance whereas the output from transistor 38 provides negative transconductance. Adjustment of the operating parameters may be obtained by adjustment of the current in the collector loads, thereby adjusting the transconductance. With such an amplifier there is a range of about 60mV in which there is a linear relation between input and output. It is therefore necessary to limit the input signal to within this range and a suitable circuit is shown as at 46 comprising a pair of transistors 48 connected in differential configuration to receive an input signal in the range of about 500mV and providing an output signal within the range 60mV. In a modification shown in Figure 5, the output load resistance R is simulated by a further amplifier 50 of similar type to amplifiers 20, 22. The output of the amplifier is connected in a feedback loop to the input and it may be shown the resistance is given by Thus by adjustment of the current controlling the transconductance, the effective load resistance may be adjusted without the need for trimming a fabricated resistance of the integrated circuit. Referring now to the oscillator section 2 shown in Figure 2 and in more detail in Figure 6, since the oscillator is in the standard Colpitts configuration, the frequency f osc of oscillation is given by:- Oscillator stability and noise performance are directly affected by the Q of the circuit, which is determined in part by the impedance of the quadrature network 4. It may be shown that in order for network 4 of Figure 2 to provide a quadrature output at the frequency of oscillation:- Referring now to Figure 6, the oscillator section 2 has a main amplifier 6 with an gain of 20db. Whilst the gyrator G1 has, as explained with reference to Figure 3, a loss factor, resulting from the used of limiting circuit 46 to limit the value of input voltage applied to transistors 36, 38 to 60mV, nevertheless it exhibits an overall gain of say 6db at resonance resulting from the Q factor of the circuit. Thus in order to provide an oscillatory circuit, it is necessary to provide a signal limiting amplifier 60 having a loss factor of say -20db, in order to provide the correct gain within the circuit. Thus the overall gain in the circuit is +6db, and this small factor ensures the circuit will always oscillate, the excess gain ensuring that amplifier 6 is drawn hard into its non-linear regions. Although the oscillator network and quadrature phase shift network have been described specifically with respect to use in combination in a Barber receiver, it will be understood that they may be used single or in combination in any type of electrical circuit for which their characteristics are suitable, and the scope of the invention is limited only by the appended claims.

Claims: A quadrature network fabricated as an integrated circuit and comprising a first capacitive means including first and second capacitors (C4, C5) connected in series between an input port of the network and a point of reference potential, and gyrator means (G2) having its input connected across said second capacitor (C5) and comprising a first amplifier (20) whose output is coupled to a second point of reference potential by way of second capacitive means (C6) and to the input of a second amplifier (22) whose output is coupled in a feedback loop to the input of the first amplifier (20), the transconductances of the first and second amplifiers (20, 22) being controllable such that a signal of a predetermined frequency applied to the input port of the quadrature network is shifted in phase by 90° at the input of the first amplifier (20), which forms the output port of the quadrature network. A sinewave oscillator circuit fabricated as an integrated circuit, the oscillator comprising a main amplifier (6) having its output coupled to its input in a feedback loop which includes a reactive network (C1, C2, G1) providing a required phase shift to sustain oscillations, the reactive network comprising a first capacitive means (C1, C2) and a gyrator (G1), the gyrator (G1) comprising a first amplifier (10) whose output is coupled to a second capacitive means (C3) and to the input of a second amplifier (12) whose output is coupled in a feedback loop to the input of the first amplifier (10), and the transconductances of the first and second amplifiers (10, 12) being controllable to control the impedance of the gyrator (G1) and thereby the frequency of oscillation of the oscillator circuit. An oscillator circuit as claimed in Claim 2 wherein the first capacitance means (C1, C2) comprises a first capacitor (C1) connected between an input of the main amplifier (6) and the input of the gyrator (G1), and a second capacitor (C2) connected between reference potential and the input of the gyrator (G1). An oscillator circuit as claimed in Claim 2 or Claim 3 wherein the first amplifier (10) is a noninverting amplifier and the second amplifier (12) is an inverting amplifier. An oscillator circuit as claimed in Claim 2, Claim 3 or Claim 4 wherein the second capacitive means comprises a capacitor (C3) connected between reference potential and the output of the first amplifier (10). An electrical circuit comprising an oscillator coupled to a quadrature phase shift network, the oscillator including a main amplifier (6) having a feedback loop including a reactive network (C1, C2, G1), the reactive network including a first capacitive means (C1, C2) and a first gyrator (G1), the first gyrator comprising a first amplifier (10) whose output is coupled to a second capacitive means (C3) and to the input of a second amplifier (12) whose output is coupled in a feedback loop to the input of the first amplifier (10), and the quadrature network including a third capacitive means (C4, C5) coupled to a second gyrator (G2), the second gyrator comprising a third amplifier (20) whose output is coupled to a fourth capacitive means (C6) and to the input of a fourth amplifier (22) whose output is coupled in a feedback loop to the input of the third amplifier (20), the arrangement being such that a signal applied at a predetermined frequency to the input of the quadrature network is shifted in phase by 90° at the output of the quadrature network, and each of said first, second, third and fourth amplifiers (10, 12, 20, 22) including means for controlling the transconductance of the amplifier, said controlling means being ganged together whereby to vary the frequency of the oscillator while maintaining a phase shift of 90° from the quadrature network at the oscillator frequency. A circuit as claimed in Claim 6 including a load resistance (R) at the network output comprising a sixth amplifier having its output connected to its input in feedback configuration. A circuit as claimed in Claim 6 wherein any of the said amplifiers (10, 12, 20, 22) comprises a pair of bipolar transistors (36, 38) whose emitters are coupled by a constant current source (40) and whose collector loads are constituted by constant current sources (42, 44), the transconductance of the amplifier (10, 12, 20, 22) being variable by adjusting the value of operating current of a collector load constant current source (42, 44). A circuit as claimed in Claim 8 wherein the input to each amplifier (10, 12, 20, 22) includes means (46) for limiting the value of the input signal to a range within which there is a linear relationship between input and output signals of the pair of bipolar transistors (36, 38).

Inventors: ATKINSON, SIMON, CARR, FRANCIS
Applicants: PLESSEY SEMICONDUCTORS LIMITED
Application year: 1988
Main classification: H04B1
IPC codes: H03B5, H03D1, H03H11, H03B27, H03D7