MULTIVIBRATOR CIRCUIT EMPLOYING FIELD EFFECT DEVICES

Abstract: A multivibrator circuit employing field effect transistors which operates with a high oscillation frequency and low power consumption comprises four series connections. The first series connection comprises a parallel connection including a resistor R₁ and a diode D₁ connected in parallel, and two field effect transistors Q₁ and Q₃, the first series connection being provided between the power supply V DD1 and the ground V SS . The second series connection comprises a field effect transistor Q₅, diodes D₃ for shifting a level and a field effect transistor Q₇, the second series connection being provided between the power supply V DD2 and the ground V SS . The third series connection comprises R₂, a diode D₂ and two field effect transistors Q₂ and Q₄ in the same manner as in the first series connection. The fourth series connection comprises a field effect device Q₆, diodes D₄ for shifting a level and a field effect device Q₈ in the same manner as in the second series connection. The first and second outputs are connected to the second and fourth series connections, respectively. A capacitor C₁ is connected between the sources of the field effect transistors Q₁ and Q₂.

Description: BACKGROUND OF THE INVENTION Field of the Invention The present invention relates to a multivibrator circuit and, more particularly, to a multivibrator circuit in which the drift of oscillation frequency with changes in ambient temperature is compensated. Description of the Prior Art Fig. 1 is a schematic diagram showing an emitter-coupled multivibrator circuit employing conventional silicon npn-type bipolar transistors. A similar circuit is described in "Bipolar and MOS Analog Integrated circuit Design" by A.B. Grebene, published by Micro-Linear Corporation in 1984. Referring to Fig. 1, the multivibrator circuit comprises four series connections between a power supply V cc and the ground V ee . The first series connection comprises a first parallel connection of a resistor R₁₁ and a diode D₁₁, and npn type bipolar transistors Q₁₁ and Q₁₃. The emitter of the transistor Q₁₁ and the collector of the transistor Q₁₃ are connected together to constitute a node N₁₃. The second series connection comprises npn type bipolar transistors Q₁₅ and Q₁₇. The emitter of the transistor Q₁₅ and the collector of the transistor Q₁₇ are connected together to constitute a node N₁₁. The node N₁₁ is connected to a first output OUT₁. The base of the transistor Q₁₅ is connected to a connection point between the collector of the transistor Q₁₁ and the first parallel connection. The third series connection comprises a second parallel connection of a resistor R₁₂ and a diode D₁₂, and npn type bipolar transistors Q₁₂ and Q₁₄. The emitter of the transistor Q₁₂ and the collector of the transistor Q₁₄ are connected together to constitute a node N₁₄. The base of the transistor Q₁₂ is connected to the first output OUT₁. A capacitor C₁₁ is connected between the node N₁₃ and the node N₁₄. The bases of the transistors Q₁₃ and Q₁₄ are connected together and a connection point therebetween is connected to a first constant voltage supply V co . The fourth series connection comprises npn type bipolar transistors Q₁₆ and Q₁₈. The emitter of the transistor Q₁₆ and the collector of the transistor Q₁₈ are connected together to constitute a node N₁₂. The node N₁₂ is connected to a second output OUT₂ . The base of the transistor Q₁₆ is connected to a connection point between the collector of the transistor Q₁₂ and the second parallel connection. The base of the transistor Q₁₁ is connected to the second output OUT₂ . The bases of the transistors Q₁₇ and Q₁₈ are connected together and the connection point therebetween is connected to a second constant supply voltage V CB . Next, description is made of operation of the circuit. Either one of the transistors Q₁₁ and Q₁₂ constituting an emitter-coupled switch is always on and the other is off. First, assuming that the transistor Q₁₁ is off and the transistor Q₁₂ is on, a current I flows from the node N₁₄ to the node N₁₃ through the capacitor C₁₁ , because the current I is always flowing through the transistors Q₁₃ and Q₁₄ serving as constant current load in response to a voltage applied to a first control input. Therefore, a current of 2I flows through the transistor Q₁₂. A collector voltage of the transistor Q₁₂ is decreased by a voltage of 2IR₂ by a resistor R₁₂ (a resistance value is also represented by R₁₂). Since a barrier voltage  B of the diode D₁₂ and a value (R₁₂) of the resistor R₁₂ are selected so as to satisfy the relation of 2IR₁₂ ≧  B' the collector voltage of the transistor Q₁₂ is clamped at V cc -  B . As a result, an emitter voltage V (N₁₂) of the transistor Q₁₆ becomes V cc - 2 B which is lower than a base voltage V cc -  B by the voltage  B . On the other hand, when an emitter voltage V (N₁₃) of the transistor Q₁₁ is reduced by the current I flowing through the capacitor C₁₁ and becomes V cc - 3 B which is lower than the base voltage V cc - 2 B of the transistor Q₁₁ by the voltage  B' the transistor Q₁₁ is turned on. As a result, a base voltage of the transistor Q₁₅ is reduced from V cc to V cc -  B and an emitter voltage V (N₁₁) of the transistor Q₁₅ is reduced from V cc -  B to V cc - 2 B . Since an emitter voltage V (N₁₄) of the transistor Q₁₂ changes with a delay of a sufficiently large time constant by the capacitor C₁₁ as compared with switching operation of the transistors Q₁₁ and Q₁₂, it is equal to an emitter voltage of V cc - 2 B when the transistor Q₁₂ is on. Therefore, the transistor Q₁₂ is turned off. After the transistor Q₁₂ is turned off, the base voltage of the transistor Q₁₁ is raised by a voltage  B and the voltages V (N₁₃) and V (N₁₄) across the capacitor C₁₁ are raised from V cc - 3 B to V cc - 2 B and from V cc - 2 B to V cc -  B' respectively. As a result, the states of the transistors Q₁₁ and Q₁₂ initially set have been changed with each other. Fig. 2 is a timing chart showing changes of the first and second output voltages V (N₁₁) and V (N₁₂) of the emitter-coupled multivibrator circuit in Fig. 1 and the voltages V (N₁₄) and V (N₁₃) of the nodes N₁₄ and N₁₃ as well as a change of the voltage difference V (N₁₄) - V (N₁₃) across the capacitor C₁₁. An oscillation frequency f₀ of the multivibrator circuit shown in Fig. 1 is represented by the following equation: f₀ = I/4 B where C is a capacitance value of the capacitor C₁₁, I is a current flowing through the transistors Q₁₃ and Q₁₄ for constant current loads of the emitter-coupled switch portion, and reference 2 B is a maximum amplitude voltage applied across the capacitor C₁₁. Since the current I can be controlled by a voltage V co applied to the bases of the transistors Q₁₃ and Q₁₄, the emitter-coupled multivibrator circuit shown in Fig. 1 is a voltage-controlled type oscillation circuit. Since the conventional emitter-coupled multivibrator circuit shown in Fig. 1 employs a silicon bipolar transistor, power consumption is high. As a consequence there is a tendency for the fluctuations in operating temperature to be large and, as a result and in the absence of compensation, the oscillating frequency can drift or become unstable. Techniques for improving temperature stability in circuits such as that shown in Fig. 1 are also described in the abovementioned article by A B Grebene. In one of these techniques the oscillating current is made proportional to the diodic voltage drop imposed on the circuit. In any of these techniques the circuit configuration is modified to eliminate voltage drop dependency in the frequency expression describing the behaviour of the circuit. In each case the details of implementation are complex. DISCLOSURE OF THE INVENTION The present invention is intended to provide a multivibrator circuit in which provision is made for compensating frequency drift or instability due to changes of temperature. A simple alternative is provided to the compensation techniques just mentioned. In accordance with the present invention there is provided a multivibrator circuit comprising: first and second alternately operable transistors; first and second current sources connected to each first and second transistor respectively; first and second voltage drop clamp circuit means, including respective first and second diode means, connected to each first and second transistor respectively; and timing capacitance means connected between a junction of said first transistor with said first current source and a junction of said second transistor with said second current source; wherein said circuit is so configured to produce an oscillating signal the frequency f₀ of which is given by the expression: f₀ = I/4 B ; where I is the current supplied from each current source, C is the capacitance of said timing capacitance means, and 2 B the range of voltage change across said capacitance means,  B being the diodic drop voltage of each of said first and second diode means; which circuit is characterized in that: each first and second current source comprises a field effect transistor and has a third diode means connected between its gate electrode and its source electrode to effect a reduction in the temperature coefficient of the oscillation frequency f₀. As just set forth, the problem of compensating temperature variation of the multivibrator oscillation frequency is solved by providing current sources of field-effect transistor (FET) construction and by connecting the gates of these transistors to their sources by means of an appropriately selected diodic shunt. The construction then is elegantly simple and is easy to implement. Since current sourcing is provided by field-effect transistors, power consumption is reduced. Where all transistors are of the field-effect type, as in the preferred embodiment described hereinafter, the power consumption is reduced considerably. At the same time it is possible to achieve stable operation at particularly high frequencies. BRIEF DESCRIPTION OF THE DRAWINGS In the accompanying drawings: Fig. 1 is a schematic diagram showing a multivibrator circuit employing conventional npn type bipolar transistors; Fig. 2 is a timing chart for explaining operation of the multivibrator circuit of Fig. 1; Fig. 3 is a schematic diagram showing a multivibrator circuit similar to that of Fig. 1 but employing field-effect transistors in place of conventional bipolar transistors; Fig. 4 is a graph showing the static characteristics of drain voltage and drain current of a field-effect transistor such as is used in the multivibrator circuit of Fig. 3; Fig. 5 is a timing chart for explaining operation of the multivibrator circuit shown in Fig. 3; Fig. 6 is a schematic diagram showing a multivibrator circuit similar to that of Fig. 3 but in which temperature compensation is provided in accordance with the present invention; Fig. 7 is cancelled; Fig. 8 is a block diagram of a high-frequency superposing circuit for reducing a return light noise of a laser diode for showing an example of application of the present invention; Fig. 9 is a schematic diagram showing an example of an amplification circuit and a laser diode drive circuit shown in the block diagram of Fig. 8. DESCRIPTION OF A PREFERRED EMBODIMENT As background to the description of Fig. 6 of the drawings (which shows the preferred embodiment of the invention) here follows a brief discussion of a field-effect transistor implementation of the Fig. 1 multivibrator circuit. Referring to Fig. 3, the multivibrator circuit employing field-effect transistors comprises four series connections. The first and third series connections are provided between a first power supply V DD1 and the ground V SS . The second and fourth connections are provided between a second power supply V DD2 and the ground V SS . The first series connection comprises a first clamp circuit in which a resistor R₁ and a diode D₁ are connected in parallel, an n type field effect transistor Q₁ constituting a source-coupled switch and an n type field effect transistor Q₃ as a current source. The transistor Q₁ has its drain connected to the first clamp circuit and its source connected to the drain of the transistor Q₃ together to constitute a node N₃. The transistor Q₃ has its source connected to the ground V SS . The second series connection comprises an n type field effect transistor Q₅ constituting a source follower, two diodes D₃ for shifting level, and an n type field effect transistor Q₇. The transistor Q₅ has its drain connected to the second power supply V DD2 and the transistor Q₅ has its source connected to the anode of the diode D₃. The drain of the transistor Q₇ and the cathode of the diode D₃ are connected together to constitute a node N₁. The node N₁ is connected to a first output OUT ₁. The transistor Q₅ has its gate connected to a connection point between the drain of the transistor Q₁ and the first clamp circuit. The transistor Q₅ and the diode D₃ constitute a first level shift circuit. The third series connection comprises a second clamp circuit in which a resistor R₂ and a diode D₂ are connected in parallel, an n type field effect transistor Q₂ constituting the source-coupled switch and an n type field effect transistor Q₄ as a current source. The transistor Q₂ has its drain connected to the second clamp circuit and the transistor Q₂ has its source connected together to the drain of the transistor Q₄ to constitute a node N₄ and its gate connected to the first output OUT₁. The transistor Q₄ has its source connected to the ground V SS . A capacitor C₁ is connected between the node N₃ and the node N₄. Gates of the transistors Q₃ and Q₄ are connected together and a connection point thereof is connected to a first constant supply voltage V CS1 . The fourth series connection comprises an n type field effect transistor Q₆ constituting a source follower, two diodes D₃ for shifting level and an n type field effect transistor Q₈. The transistor Q₆ has its drain connected to the second power supply V DD2 and the transistor Q₆ has its source connected to the anode of the diode D₄ The transistor Q₆ has its gate connected to a connection point of the drain of the transistor Q₂ and the second clamp circuit. The transistor Q₆ and the diode D₆ constitute the second level shift circuit. The transistor Q₈ has its drain connected together to be cathode of the diode D₄ to constitute a node N₂. The node N₂ is connected to a second output OUT₂ . The transistor Q₁ has its gate connected to the second output. The gates of the transistors Q₇ and Q₈ are connected together and a connection point thereof is connected to a second constant supply voltage V CS2 . The transistors Q₃ and Q₄ are constant power supply and control an amount of current flowing through the transistors Q₃ and Q₄, respectively, in response to the first constant supply voltage V CS1 . The transistors Q₇ and Q₈ are also constant power supplies and they control an amount of current flowing through the transistors Q₇ and Q₈, respectively, in response to a voltage applied to the second supply voltage V CS2 . Resistors R₁ and R₂ are load resistances of the transistors Q₁ and Q₂, respectively (the resistance values are R₁ and R₂, respectively). The diodes D₁ and D₂ limit a voltage drop due to the resistors R₁ and R₂, respectively. Both barrier voltages of the diodes D₁ and D₂ are represented by  B . In general, static characteristics of a field effect transistor are represented by the following equations: where I D is the drain current of the field effect transistor, V D is the drain voltage, V G is the gate voltage, V TH is a threshold voltage of the gate and K is a transconductance. Fig. 4 is a graph showing static characteristics of the field effect device represented by the equation (1). Referring to Fig. 4, the field effect device performs constant current operation in a saturation region II shown in Fig. 4 regardless of the voltage applied to the drain. Referring to the field effect transistor multivibrator circuit shown in Fig. 3, it is assumed that the transistors Q₁, Q₂, Q₅, and Q₆ operate in a saturation region. Assuming that gate voltages of the transistors Q₁, Q₂, Q₅ and Q₆ are threshold voltage V TH when a current I₁ flowing through the transistors Q₃, Q₄, Q₇ and Q₈, respectively serving as constant current loads is equal to the respective drain currents of the transistors Q₁, Q₂, Q₅ and Q₆, the respective source voltages of the transistors Q₁ and Q₂ are reduced by V TH as compared with the gate voltages when the transistors Q₁ and Q₂ are on. Respective source voltages of the transistors Q₅ and Q₆ are reduced by V TH as compared with the gate voltages. Now, description is made of the operation of the transistors Q₁, Q₂, Q₅ and Q₆ in the saturation region. Assuming that Q₁ is off and Q₂ is on, the respective source, gate and drain voltages of the transistors Q₁, Q₂, Q₅ and Q₆ are shown in table 1. The field effect transistors operates in the saturation region on conditions that V G - V TH < VD from the above described equation (1) is satisfied. Therefore, when the condition of which solution of inequality has been found from the table 1, is satisfied, the transistors Q₁, Q₂, Q₅ and Q₆ operate in the saturation region. source voltage gate voltage drain voltage Q₅ (ON) V DD1 - V TH V DD1 V DD2 Q₂ (ON) V DD1 - 2 B - 2V TH V DD1 - 2 B - V TH V DD1 -  B Q₆ (ON) V DD1 -  B - V TH V DD1 -  B V DD2 Q₁ (OFF) V DD1 - 3 B - 2V TH V DD1 - 3 B - V TH V DD1 V DD1 -  B - 2V TH When the transistors Q₁, Q₂, Q₅ and Q₆ can operate in the saturation region by increasing the number of the diodes D₃ and D₄ in Fig. 3 to three or more. On the other hand, in the multivibrator circuit shown in Fig. 3, assuming that the relation between  B and V TH is the transistor Q₂ operates in a non-saturation region and the transistors Q₁, Q₅ and Q₆ operate in the saturation region, as it is obvious from the table 1. The voltage between the gate and source of the transistor Q₂ is reduced by more than V TH and a maximum voltage amplitude across the capacitor C₁ is less than 2 B and dependent on the V TH . Therefore, a high oscillation frequency is provided and it is more difficult to control the threshold voltage V TH than to control  B . As a result, an oscillation frequency cannot be provided with good repeatability. Description is now made of the operation of the multivibrator circuit in Fig. 3. Either one of the transistors Q₁ and Q₂ constituting the source-coupled switch is on and the other is off. First, assuming that the transistor Q₁ is off and the transistor Q₂ is on, a current I flows from the node N₄ to the node N₃ through the capacitor C₁ because the current I always flows through the respective transistors Q₃ and Q₄ serving as the current sources. Therefore, a current of 2I flows through the transistor Q₂. Although a drain voltage of the transistor Q₂ is reduced by 2IR₂ by the resistor R₂, the drain voltage of the transistor Q₂ is clamped at V DD1 -  B if the value of the resistor R₂ is selected as a barrier voltage  B of the diode D₂ so as to satisfy the relation of 2IR₂ ≧  B . As a result, the source voltage of the transistor Q₆ becomes a voltage of V DD1 -  B - V TH which is lower than a gate voltage of V DD1 -  B by the threshold voltage V TH of the transistor Q₆. At this time, the level of the gate voltage V (N₂) of the transistor Q₁ is shifted to V DD1 - 3 B - V TH by a level shift diode D₄. On the other hand, the source voltage V (N₃) of the transistor Q₁ is reduced by the current I flowing through the capacitor C₁. When the source voltage of the transistor Q₁ becomes a voltage of VDD1 - 3 B -2V TH which is lower than the gate voltage V DD1 - 3 B - V TH of the transistor Q₁ by the threshold voltage V TH of the field effect transistor Q₁, the transistor Q₁ is turned on. The gate voltage of the transistor Q₅ is reduced from V DD1 to V DD1 -  B and the source voltage of the transistor Q₅ is reduced from V DD1 - V TH to V DD1 - B - V TH . The gate voltage V (N₁) of the transistor Q₂ is reduced from V DD1 - 2 B - V TH to V DD1 - 3 B - V TH by the level shift diode D₃. Since the source voltage (N₄) of the transistor Q₂ is clamped at sufficiently large time constant by the capacitor C₁ as compared with the switching time of two transistors, it is equal to the source voltage V DD1 - 2 B - 2V TH when the transistor Q₂ is on. Therefore, the transistor Q₂ is turned off. When the transistor Q₂ is turned off, the gate voltage of the transistor Q₁ is raised by  B and the voltage V (N₃) and V (N₄) across the capacitor C₁ are raised from V DD1 - 3 B - 2V TH to V DD1 - 2 B - 2V TH and from V DD1 - 2 B - 2V TH to V DD1 -  B - 2V TH , respectively. As a result, the initial states in which the transistor Q₁ was off and the transistor Q₂ was on have changed with each other. Fig. 5 is a timing chart showing changes in the voltages V (N₁) and V (N₂) of the first and second outputs, the potentials V (N₄) and V (N₃) across the capacitor C₁ and the voltage V (N₄) - V (N₃) across the capacitor in the multivibrator circuit shown in Fig. 3. An oscillation threshold frequency f₀ is determined by capacitance C of the capacitor C₁, a current value I flowing through the transistors Q₃ and Q₄ serving as power supply of the source-coupled switch portion and a maximum amplitude voltage 2 B across the capacitor C₁ and the following equation is obtained: In the field effect transistor multivibrator circuit shown in Fig. 3, the oscillation frequency f₀ becomes higher because the oscillation frequency f₀ depends on the barrier voltage  B when the ambient temperature becomes higher, and the barrier voltage  B is reduced. The below described improvements will effectively prevent the ambient temperature from affecting the oscillation frequency f₀. Fig. 6 is a schematic diagram showing another embodiment of a multivibrator circuit employing field effect transistors in accordance with the present invention. The multivibrator circuit in Fig. 6 is additionally provided with a bias resistor R₃ connected between a first constant supply voltage V CS1 and a connection point of gates of the transistors Q₃ and Q₄, and a diode D₅ for temperature compensation connected between the connection point of the gates of the transistors Q₃ and Q₄ and the ground V SS , as compared with the multivibrator circuit in Fig. 3. Although only one diode D₅ is shown in Fig. 6, a series connection of one or more diodes may also be applicable. Since other circuit portions of Fig. 6 are the same as those in Fig. 3, a description thereof is omitted. Next, a description is made of operation of the multivibrator circuit of Fig. 6. An oscillation frequency f₀ of the multivibrator circuit in Fig. 6 is also represented by the equation (2), which is the same as that of the circuit in Fig. 3. When the ambient temperature is raised during operation, the barrier voltage  B of the diodes D₁ and D₂ is lowered. However, since the barrier voltage  B of the diode D₅ is also lowered at the same time, the gate voltages of the transistors Q₃ and Q₄ are lowered and the value I of the drain current flowing through the transistors Q₃ and Q₄ are also reduced. Therefore, since the rise of the oscillation frequency due to the drop of the barrier voltage  B can be offset by the decrease of the drain current value I, the oscillation frequency f₀ depends less on the temperature. A temperature coefficient of the barrier voltage  B is generally represented by and the drain current value I in the saturation region of a field effect transistor is represented by the following equation: I = K (V g - V TH )² (3) where V g is a gate voltage, V TH is a threshold voltage and K is a coefficient. Referring to Fig. 6, since the gate voltages V g Of the transistors Q₃ and Q₄ are equal to the barrier voltage  B , the equation (3) is also represented by the following equation: I = K ( B - V TH )² (4) Assuming that both of coefficient K and the temperature coefficient of the threshold voltage V TH are significantly smaller than the temperature coefficient of the barrier voltage  B, a temperature coefficient of the drain current I is represented by the following equation: and a temperature coefficient of the oscillation frequency f₀ is represented by the following equation: Therefore, if the threshold voltage V TH and the barrier voltage  B is selected so as to satisfy the relation of V TH = - B , the temperature coefficient of the oscillation frequency f₀ can be reduced to the value which is a product of and the temperature coefficient of the oscillation frequency f₀ of the multivibrator circuit shown in Fig. 3. It is understood that the number of diodes D₁, D₂, D₃ and D₄ of the multivibrator circuit in accordance with present invention shown in Fig. 6 is not limited to the number of diodes shown in the figure and series connection of one or more diodes can be applied. A description is made of an example of application of the present invention in the following. With recent developments in optical technologies, a demand for an oscillator having high operating speed and outputting an output signal with a high frequency in a stable manner has been increased. In an optical disc, optical fiber communication, or optical fiber gyroscope, for example, a high frequency signal is applied to a laser diode using a high frequency superposed circuit in order to reduce light induced noise due to return light. A single mode laser of a refractive index waveguide type oscillates in a multi-mode so that noise due to return light can be effectively reduced. Some reports as indicated below are made with regard to this effect. (1) Epworth, R E, "The Phenomenon of Modal Noise in Analog and Digital Optical Fiber Systems", Proc. 4th. Europ. Conf. Opt. Commun. (Genova, Italy), pp 492-501, Sept 1978. (2) Vanderwall, J and Blackburn, J, "Suppression of Some Artifacts of Modal Noise in Fiber-Optic Systems", Opt. Lett Vol. 4, No 9, pp 295-296, Sept 1979. (3) Sato, K and Asatani, K, "Speckle Noise Reduction in Fiber Optic Analog Video Transmission Using Semiconductor Laser Diode", IEEE Tr. Commun., Vol. COM-29, No 7, pp 1017- 1024, July 1981. Fig. 8 is a block diagram of a high frequency superposed circuit for reducing the return light noise of a laser diode as an example of an application of the present invention. Referring to Fig. 8, the high frequency superposed circuit comprises a high frequency multivibrator circuit 11, an amplitude circuit 12 amplifying a signal generated from the circuit 11 and a laser diode drive circuit 13. A laser diode 14 is connected to an output of the laser diode drive circuit 13 and an APC (Automatic Power control) circuit 15 for adjusting an optical output of the laser diode is also connected to the laser diode drive circuit 13. Fig. 9 is a schematic diagram showing one concrete example of the amplitude circuit 12 and the laser diode drive circuit 13 shown in Fig. 8. Referring to Fig. 9, the amplitude circuit 12 comprises a circuit of a first stage connected between a power supply V D1 and the ground V SS and a circuit of a second stage connected between a power supply V D2 and the ground V SS . The circuit of the first stage comprises an n channel field effect transistor 65 connected to a parallel connection which is formed between a series connection of a load 61 and an n channel field effect transistor 62 and a series connection of a load 63 and an n channel field effect transistor 64. The transistor 62 has its gate connected to an input IN₁ and the transistor 64 has its gate connected to an input IN₁ . The inputs IN₁ and IN₁ are connected to receive signals of outputs OUT₁ and OUT₁ of the multivibrator circuit of the present invention, respectively. The transistor 65 has its gate connected to a constant supply voltage V C1 . The circuit of the second stage comprises an n channel field effect transistor 67, two diodes 68 for level shifting and an n channel field effect transistor 69 connected in series between the power supply V D2 and the ground V SS . It further comprises an n channel field effect transistor 70, two diodes 71 for level shifting and an n channel field effect transistor 72 connected in series between the power supply V D2 and the ground V SS . The transistor 67 has its gate connected to the connection point between the load 61 and the transistor 62. The transistor 70 has its gate connected to the connection point between the load 63 and the transistor 64. The gate of the transistor 69 and the gate of the transistor 72 are connected together to a constant supply voltage V C2 . In the laser diode drive circuit 13, a load 73 and an n channel field effect transistors 74 and 76 are connected in series between a power supply V D3 and the ground V SS . The transistor 74 has its gate connected to the connection point between the diodes 68 and the transistor 69. The transistor 76 has its gate connected to a constant supply voltage V C3 . A laser diode 14 and an n channel field effect transistor 77 are connected in series between a power supply V D4 and the ground V SS . The transistor 77 has its gate connected to an input IN₂. The input IN₂ is connected to receive an output signal of an APC circuit 15. An n channel field effect transistor 75 is connected between the connection point between the laser diode 14 and the transistor 77 and the connection point between the transistors 74 and 76. The transistor 75 has its gate connected to the connection point between the diode 71 and the transistor 72. In operation of the high frequency superposed circuit shown in Fig. 9, the multivibrator circuit generates high frequency signal (several hundred MHz∼ several GHz) necessary to reduce the light induced noise due to return light of the laser diode and applies the same to the inputs IN₁ and IN₁ of the amplitude circuit 12. The amplitude circuit 12 receives the high frequency signal and the laser diode amplifies this signal to a signal level necessary for oscillation in a multimode. Then, the laser diode drive circuit 13 receives the amplified high frequency signal to apply it to the laser diode 14. The laser diode drive circuit 13 also adjust a DC bias current flowing through the laser diode 14 in response to an output signal of the APC circuit 15 applied to the input IN₂. In addition, the amplitude circuit 12 may have an AGC (Automatic Gain Control) function to make an optimum adjustment of the output level of the high frequency superposed signal. As described above, by applying the multivibrator circuit to the high frequency superposed circuit shown in Fig. 8, the following advantages are brought about. (1) A high frequency signal which is necessary to reduce the light induced noise due to return light of a laser diode can be readily obtained. (2) A monolithic integration can be easily attained because a circuit can be formed using the same kind of compound semiconductors as that of laser diodes, e.g. GaAs field effect transistors. (3) By employing a source-coupled type circuit, it is made easy to connect the same type, that is, source-coupled type amplitude circuit and laser diode drive circuit as shown in Fig. 9, for example. (4) By employing a source-coupled type circuit, the circuit can be driven by a single power supply. Although the present invention has been described and illustrated in detail, it is clearly to be understood that the same is by way of illustration and example only and is not to be taken by way of limitation, the scope of the present invention being limited only by the terms of the appended claims.

Claims: A multivibrator circuit (fig. 6) comprising: first and second alternately operable transistors (Q₁, Q₂); first and second current sources (Q₃, Q₄) connected to each first and second transistor (Q₁,Q₂) respectively; first and second voltage drop clamp circuit means (R₁, D₁ and R₂, D₂), including respective first and second diode means (D₁, D₂), connected to each first and second transistor (Q₁, Q₂) respectively; and timing capacitance means (C₁) connected between a junction (N₃) of said first transistor (Q₁) with said first current source (Q₃) and a junction (N₄) of said second transistor (Q₂) with said second current source (Q₄) wherein said circuit is so configured to produce an oscillating signal the frequency f₀ of which is given by the expression: f₀ = I/4C B ; where I is the current supplied from each current source, (Q₃,Q₄), C is the capacitance of said timing capacitance means (C₁), and 2 B is the range of voltage change across said capacitance means (C₁),  B being the diodic drop voltage of each of said first and second diode means (D₁,D₂); which circuit is characterized in that: each said first and second current source (Q 3, Q₄) comprises a field effect transistor (Q₃,Q₄) and has a third diode means (D₅) connected between its gate electrode and its source electrode to effect a reduction in the temperature coefficient of the oscillation frequency f₀. A multivibrator circuit as claimed in claim 1 wherein the gate electrodes of each of said field effect transistors (Q₃,Q₄) are connected together (N₅), the source electrodes of each of said field effect transistors are connected together (V SS ), and a common diode means (D₅) is provided as said third diode means. A multivibrator circuit as claimed in either one of the preceding claims wherein the gate electrode of each of said field effect transistors (Q₃,Q₄) is connectable to a source of bias voltage (V cs1 ) via a coupling resistance means (R₃) provided therefor. A multivibrator circuit as claimed in any one of the preceding claims wherein said first and second alternately operable transistors (Q₁,Q₂) also are field effect transistors. A multivibrator circuit as claimed in claim 3 or 4 further comprising: a first power supply rail ("V DD1 ") connected to said first and second voltage drop clamp circuit means (R₁, D₁ & R₂,D₂); first and second drive transistors (Q₅, Q₆) each of field effect transistor type and having a gate electrode connected to said first and second voltage drop clamp circuit means (R₁, D₁ & R₂, D₂) respectively; first and second level-shifting means (D₃,D₄), connected to the source electrode of each of said first and second drive transistors (Q₅,Q₆) respectively, and cross-coupled at their opposite extremity (N₁,N₂) to the gate electrode of each of said second and first alternately operable transistors (Q₂, Q₁) respectively so that the latter are each operable in saturation; and; third and fourth current supply means (Q₇,Q₈) connected to said first and second level-shifting means (D₃,D₄). A multivibrator circuit, as claimed in claim 5, comprising a second power supply rail ("V DD2 ") connected to the drain electrode of each of said first and second drive transistors (Q₅,Q₆). A multivibrator circuit, as claimed in claim 6, wherein said first and second power supply rails ("V DD1 ", "V DD2 ") are connected to lower and higher voltage power supply sources (V DD1 , V DD2 ) respectively whereby said first and second drive transistors (Q₅, Q₆) also are each operable in saturation. A multivibrator circuit, as claimed in any one of the preceding claims wherein each field effect transistor is of gallium arsenide metal-semiconductor (GaAs MESFET) type.

Inventors: TANINO, NORIYUKI MITSUBISHI DENKI K.K.LSI
Applicants: MITSUBISHI DENKI KABUSHIKI KAISHA
Application year: 1988
Main classification: H03K3
IPC codes: H03K3