A divider circuit arrangement and a dual branch receiver having such a divider circuit arrangement

Abstract: A divider circuit arrangement in which in order to avoid dividing by zero the divisor (V d ) is modified by the addition of an extra signal (X a ) to form a modified divisor V′ d = V d + X a and the dividend (V i ) is modified by the addition of the product of the quotient (V o ) and the extra signal (X a ) to form a modified dividend V′ i = V i + V o X a . A particular but not exclusive application of this divider circuit arrangement is in normalising an output signal from a dual branch receiver (not shown).

Description: The present invention relates to a divider circuit arrangement and particularly, but not exclusively, to a dual branch receiver having such a divider circuit arrangement. In analogue signal processing, a division function is often used for normalising signal amplitudes. One disadvantage associated with the division function is the possibility that the resulting quotient will go to infinity if the divisor becomes zero. When this occurs the circuit that performs this normalisation function will swing to its extreme state, for example, saturation of an analogue circuit. In practice, special precautions are usually taken to avoid this possible overflow state. European Patent Specification 0075707B1 discloses a ring interferometer in which non-zero divide by zero is avoided. Light from a laser is arranged to pass through two spatially separated, partially transmitting mirrors. Two opto-electronic sensors detect light reflected by these mirrors. The outputs from these sensors are coupled to quotient forming means. In order to avoid a "divide by zero" problem an output of one of the sensors forms the dividend and the divisor is formed by the sum of proportionate parts of the signals appearing at the outputs of the sensors. In the field of telecommunications, for example in a dual branch receiver or demodulator of a type disclosed by J.K. Goatcher, M.W. Neale and I.A.W. Vance in an article entitled "Noise considerations in an integrated circuit VHF radio receiver" in the Proceedings of the IERE Clerk Maxwell Commemorative Conference on Radio Receivers and Associated Systems (IERE Proceedings 50), University of Leeds, 7th to 9th July 1981, pages 49 to 51 a signal is normalised by it being divided using a divisor formed by the sum of the squares of the in-band components of the quadrature related signals which have been produced by mixing an input signal down to baseband. If the input signal is lost due to say a fade which may occur in a mobile environment then a divide by zero situation occurs. If such a situation should occur frequently then an inpleasant audio output may occur. An object of the present invention is to avoid a divide by zero situation arising. According to the present invention as set forth in claim 1 there is provided a divider circuit arrangement in which in order to avoid dividing by zero the divisor is modified by the addition of an extra signal and the dividend is modified by combining it with the product of the quotient and the extra signal. An embodiment of the present invention provides a divider circuit arrangement in which a first signal is to be divided by a second signal, comprising a divider having a first input for a dividend, a second input for a divisor and an output, summing means having a first input for the second signal, a second input for an extra signal and an output for the sum of the second signal and the extra signal which forms the divisor which is applied to the second input of the divider, multiplying means having a first input connected to the output of the divider, a second input connected to receive the extra signal and an output, and signal combining means having a first input for the first signal, a second input connected to receive the product signal from the multiplying means and an output for providing the desired combination of the first signal and said product signal which combination forms the dividend and is applied to the first input of the divider. The invention is based on the recognition of the fact that if the divisor (V d ) is modified by the addition of an extra signal (X a ) then the modified divisor (V' d ) will not become zero, thus V' d = V d + X a . However it is then necessary to remove the effect of the extra signal (X a ) from the final output (V o ). In accordance with the present invention this is achieved by multiplying the output (V o ) by the extra signal (X a ) and forming a combination, for example the sum of the product (V o .X a ) and the dividend signal (V i ) to form a modified dividend (V' i ), thus V' i = V i + V o X a and Since this result corresponds to the original dividend being divided by the original divisor then the influence on the final output due to adding the extra term(X a ) to the divisor (V d ) has now been totally removed. The choice of X a could be either a constant value or any function which will not allow the absolute value of V′ d from becoming zero. In making this choice of the added term X a account has to be taken of the nature of V d , that is whether it is unipolar or bipolar. For reasons of stability and dynamic range, the value of X a should be kept to the minimum, that is, it should be a small fraction of the desired output V o . If desired the value of X a could be made adaptive in response to the signal level. In an embodiment of the present invention in which the second, divisor, signal (V d ) is unipolar and the extra signal has the same polarity then the output of the multiplying means comprises a negative feedback signal to the second input of the signal combining means which is operative to form the difference between the first signal and said product signal. The first input of the summing means comprises means for multiplying the second signal by -1 and the second input to the summing means is an inverting input for inverting the extra signal. In another embodiment of the present invention the second signal is bipolar and signal transforming means are connected to the first input of the summing means for transforming the bipolar second signal into a unipolar signal. In one version of this embodiment the signal transforming means comprises a squaring circuit and a second multiplying means is provided which has its output connected to the first input of the signal combining means which in this embodiment functions as an adder, a first input of the second multiplying means being connected to receive the first signal and the second input of the second multiplying means being connected to receive the second signal. In another version of this embodiment, the signal transforming means comprises a squaring circuit and a second multiplying means is provided. The second multiplying means has a first input connected to the output of the divider, a second input connected to receive the second signal and an output for the quotient of the first signal divided by the second signal. In a further embodiment of the present invention in which the extra signal is adaptive, the divider circuit arrangement further comprises a squaring circuit coupled to the output of the divider and means for providing an output signal which comprises a substantially fixed fraction of the signal applied to its input which is coupled to an output of the squaring circuit, said output signal constituting said extra signal. Signal clamping means may be connected between the output of the signal combining means and the first input of the divider. The signal clamping means serves to limit the dynamic range of the numerator input and enable the divider to operate within its linear region thus avoiding circuit saturation and latch-up problems. The present invention further provides a dual branch receiver comprising an input for an input signal to be demodulated, quadrature related mixing means for frequency down converting the input signal to form quadrature related first and second signals, filtering means for providing in-band components of the first and second signals, first and second multiplying means, said first multiplying means forming the product of the differential with respect to time of the in-band components of the first signal multiplied by the in-band components of the second signal, said second multiplying means forming the product of the differential with respect to time of the in-band components of the second signal multipled by the in-band components of the first signal, means for subtracting the output signal produced by one of the first and second multipliers from the output signal produced by the other of the first and second multipliers and signal normalising means connected to an output of the subtracting means, said signal normalising means comprising the divider circuit arrangement made in accordance with the present invention, said first signal being derived from an output of the subtracting means and said second signal comprising the sum of the squares of the in-band components of the first and second signals obtained from the filtering means. If desired d.c. blocking capacitors may be provided in the signal paths from the quadrature related mixing means, for example in the output circuits of the filtering means. The present invention will now be described, by way of example, with reference to the accompanying drawings, wherein: Figure 1 is a block schematic diagram of a first embodiment of the present invention, Figure 2 is a block schematic diagram of a second embodiment of the present invention having a unipolar divisor, Figures 3 and 4 are block schematic diagrams of third and fourth embodiment of the present invention having a bipolar divisor, Figure 5 is a block schematic diagram of a fifth embodiment of the present invention in which the extra signal (X a ) is made adaptive, and Figure 6 is a block schematic diagram of an embodiment of a dual branch receiver having a divider circuit arrangement made in accordance with the present invention. In the drawings corresponding features have been referenced using the same reference numerals. In a divider or normalising circuit arrangement an input signal V i constituting the dividend is divided by another signal V d constituting the divisor to provide a quotient in the form of an output signal V o . Thus V o = V i / V d (1) Now if the divisor V d becomes zero, the divider circuit arrangement will saturate at its extreme state. V d can become zero if it is oscillating between positive and negative values, for example a sinusoid function, or if V d is a function of V i and V i becomes zero. Figure 1 illustrates one embodiment of a divider circuit arrangement in which measures are taken to avoid V d becoming zero. In essence an extra signal X a is added to V d in a summing circuit 10 to form a modified divisor V′ d which is applied to a divider 12. Thus V′ d = V d + X a (2) The choice of X a could be either a constant value or any function which will not allow the absolute value of V′ d to become zero. Having provided a modified divisor V′ d it is also necessary to remove the effect of the extra signal X a from the final output V o of the illustrated divider circuit arrangement. In the illustrated embodiment the output V o is fed back to a multiplier circuit 14 in which it is multiplied by the extra signal X a and the product V o X a is combined, in this embodiment added, with the input signal V i in a signal combining circuit 16 to form a modified dividend V′ i where V′ i = V i + V o X a (3) The quotient V o = V′ i /V′ d (4) Substituting equations (2) and (3) into equation (4) yields As equation(5) gives exactly the same result as equation (1), then it confirms that the influence on the final output due to adding the extra signal X a to the divisor V d has now been totally removed, whilst at the same time a divide-by-zero problem has been avoided. Subject to the foregoing comments on the choice of the extra signal X a , another factor to be taken into account is the nature of V d , that is whether it is unipolar or bipolar. Additionally for practical purposes of stability and dynamic range, the value of X a should be kept to the minimum, that is, it should be a small fraction of the desired output, V o . The value of X a can be made adapative in response to the signal level and an adaptive embodiment will be described later with reference to Figure 5 of the accompanying drawings. If the divisor V d is unipolar, that is, V d ≦ 0 or V d ≧ 0 then a non-zero positive X a should be adopted when V d ≧ 0 and a non-negative X a is adopted for V d ≦ 0. Figure 1 shows the feedback term being added to the input signal V i in the signal combining circuit 16 for V d ≧0. When a negative feedback term is desirable then this can be done using the embodiment shown in Figure 2. The signal combining stage 16 forms the difference between V i and the negatively fed back signal V o X a . In the case of a positive going divisor V d then its polarity is changed by multiplying by -1 in a multiplying circuit 18 and it is added to -X a in the summing circuit 10 so that V′ d = -(V d +X a ) . The remainder of this embodiment is the same as described with reference to Figure 1. The desirable output is now given by where V d ≧ 0 and X a is non-zero and positive. Figures 3 and 4 are embodiments in which the divisor V d is bipolar of oscillatory and can go to zero. In order to keep X a small, an option such as X a being constant and having an absolute value greater than the absolute peak value of V d is not viable. The embodiments of Figures 3 and 4 avoid the problem of a bipolar V d by transforming it into a unipolar signal by squaring the bipolar V d in a multiplier 20. The squared value of V d , that is V d ², is added to the extra signal X a in the summing circuit 10 to form a modified divisor V′ d = V d ² + X a . In Figure 3 the influence of the squaring of V d on the final output V o is cancelled by multiplying V i with V d using a multiplying circuit 22 and the product is added to V o X a in the signal combining circuit 16. The final output V o is given by The alternative technique shown in Figure 4 for avoiding the influence of squaring V d is to produce an intermediate output V′ o and multiply it by V d in a multiplying circuit so that the final output becomes Figure 5 is an adaptive embodiment of the divider circuit arrangement in which the extra signal X a is a function of the desired output V o and is given by X a = |V o | x KFB where KFB is a gain constant. In order to obtain a unipolar signal |V o | the output is applied to a full wave rectifier 26. The resultant signal is multiplied by the gain constant KFB produced by a circuit 28 which can be implemented as a resistive attenuation network. A delay network 30 is provided between the output V o and the full wave rectifier 26 to compensate for signal propagation delays. However the delay network 30 may not be needed if the signal propagation delay introduced by the circuits in the feedback loop formed by circuits 26, 28 and 14 are sufficient. The modified dividend signal V′ i is given by V′ i = V i + V o X a . A clamping circuit 32 is connected between the signal combining circuit 16 and the divider 12 to limit the value of V′ i within a range -A to +A. By limiting the dynamic range of the modified dividend V′ i , the maximum allowable output signal value V o and input signal value V i are, respectively, given as follows: The maximum value of V o occurs when V d is zero. From equations (10) and (11), it can be observed that the maximum values of V i and V o are governed by the values of input clamping level A and the feedback factor KFB. By a proper choice of these two values the divider 12 will operate within its linear region, thus avoiding circuit saturation and latch up problems. If a very small feedback factor KFB, for example 0.01 is adopted, then the reduction in dynamic range of V i will be minimal. The divider circuit arrangements shown in Figures 1 to 5, can be used in any suitable desired practical application. One example is illustrated in Figure 6 which shows a dual branch radio receiver in which a demodulated signal is normalised using a divider circuit arrangement made in accordance with the present invention. The illustrated receiver circuit is in many respects known in the art, for example the article "Noise considerations in an integrated circuit VHF radio receiver" by J.K. Goatcher, M.W. Neale and I.A.W. Vance referred to in detail in the preamble. For the sake of completeness the circuit will be described briefly. An incoming signal angle modulated on a nominal carrier frequency f c is received by an antenna 34 and is coupled by way of a band-pass anti-harmonic filter 36 to inputs of first and second quadrature related mixers 40. A local oscillator 42 of substantially the same frequency as the carrier frequency f c is applied to the mixer 38 and via a 90° phase shifter 44 is applied to the mixer 40. The outputs from the mixers 38, 40 respectively comprise in-phase signal components I and quadrature phase signal components Q at baseband frequencies. Low pass filters 46, 48 pass the in-band signal components of the I and Q signals, respectively. D.C. blocking filters 50, 52 are connected to the filters 46, 48 in order to eliminate the d.c. offsets in the filtered I and Q signals, which offsets may exceed the amplitude of the wanted signal. The in-band components of the I and Q signals from the blocking filters are differentiated with respect to time in differentiating circuits 54, 46 and are applied to respective first inputs of mixers 58, 60. These I and Q signals are applied to second inputs of the mixers 60, 58, respectively. An output of one of the mixers 58, 60 is subtracted from the output of the other of the mixers 58, 60 in a subtracting stage 62. As is known the signal at the output of the subtracting stage 62 has a square-law dependence on the level of the input signal. In order to remove this dependence, the signal at the output of the subtracting stage 62 is normalised using an amplitude divider. The divisor is obtained by summing the squares of the I and Q signals using multipliers 64, 66 and summing stage 68. In a situation of a zero input signal due to say a fade then the divisor will become zero leading to saturation and latching-up in the divider. This problem is resolved by providing the divider circuit arrangement made in accordance with the present invention and connecting the arrangement so that its input signal V i is the output of the subtracting stage and V d is the sum of the squares of the I and Q signals at the output of the summing stage 68. By avoiding the risk of dividing by zero then the likelihood of an unpleasant audio output being produced is slight. The demodulated signal is obtained from the output of a low pass filter 70 connected to the divider 12. The divider circuit arrangements illustrated in Figures 2 to 5 can be used in place of the arrangement illustrated which is essentially that of Figure 1.

Claims: A divider circuit arrangement in which in order to avoid dividing by zero the divisor input signal (V d ) is modified by the addition of an extra signal (X a ) and the dividend input signal (V i ), is modified by combining it with a feedback signal which is the product of the quotient and the extra signal in such a manner that the influence of the extra signal is eliminated in the output signal (V o ) representing the quotient. A divider circuit arrangement according to claim 1 wherein in which a first signal is to be divided by a second signal, comprising a divider (12) having a first input for a dividend (V i ), a second input for a divisor (V' d ) and an output (V o ), summing means (10) having a first input for the second signal (V d ), a second input for an extra signal (X a ) and an output for the sum of the second signal and the extra signal which forms the divisor which is applied to the second input of the divider, multiplying means (14) having a first input connected to the output of the divider, a second input connected to receive the extra signal and an output, and signal combining means (16) having a first input for the first signal, a second input connected to receive the product signal from the multiplying means and an output for providing the desired combination of the first signal and said product signal which combination forms the dividend and is applied to the first input of the divider. An arrangement as claimed in claim 2, wherein when the second signal is unipolar, the extra signal has the same polarity as the second signal. An arrangement as claimed in claim 3, wherein the output of the multiplying means (14) comprises a negative feedback signal to the second input of the signal combining (16) means which is operative to form the difference between the first signal and said product signal, the first input of the summing means (10) comprises means for multiplying the second signal by -1 and the second input to the summing means is an inverting input. An arrangement as claimed in claim 2, wherein the second signal is bipolar and signal transforming means (20) are connected to the first input of the summing means (10) for transforming the bipolar second signal into a unipolar signal. An arrangement as claimed in claim 5, wherein said signal transforming means comprises a squaring circuit (20) and wherein a second multiplying means (22) is provided which has its output connected to the first input of the signal combining means, a first input of the second multiplying means being connected to receive the first signal and the second input of the second multiplying means being connected to receive the second signal. An arrangement as claimed in claim 5, wherein said signal transforming means comprises a squaring circuit (20), and wherein a second multiplying means (24) is provided, the second multiplying means having a first input connected to the output of the divider (12), a second input connected to receive the second signal and an output for the quotient of the first signal divided by the second signal. An arrangement as claimed in claim 2, wherein the extra signal is adaptive and comprises a fraction of the output signal from the divider. An arrangement as claimed in claim 8, wherein the extra signal is unipolar and the arrangement further comprises a squaring circuit (26) coupled to the output of the divider and means (28) for providing an output signal which comprises a substantially fixed fraction of the signal applied to its input which is coupled to an output of the squaring circuit, said output signal constituting said extra signal. An arrangement as claimed in claim 9, wherein signal clamping means are connected between the output of the signal combining means and the first input of the divider. A dual branch receiver comprising an input for an input signal to be demodulated, quadrature related mixing means (40) for frequency down converting the input signal to form quadrature related first (I) and second signals, filtering means for providing in-band components of the first and second (Q) signals, first (58) and second (60) multiplying means, said first multiplying means forming the product of the differential with respect to time of the in-band components of the first signal multiplied by the in-band components of the second signal, said second multiplying means forming the product of the differential with respect to time of the in-band components of the second signal multiplied by the in-band components of the first signal, means (62) for subtracting the output signal produced by one of the first and second multipliers from the output signal produced by the other of the first and second multipliers, and signal normalising means connected to an output of the subtracting means, said signal normalising means comprising the divider circuit arrangement as claimed in any one of claims 1 to 10, said first signal being derived from an output of the subtracting means and said second signal comprising the sum of the squares of the in-band components of the first and second signals obtained from the filtering means. A receiver as claimed in claims 11, further comprising d.c. blocking capacitors (50, 52) in the signal paths from the quadrature related mixing means. A receiver as claimed in claim 12, wherein the d.c. blocking capacitors are provided in output circuits of the filtering means.

Inventors: CHUNG KAH-SENG, CHUNG, KAH-SENG
Applicants: PHILIPS NV, N.V. PHILIPS' GLOEILAMPENFABRIEKEN
Application year: 1988
Main classification: G06G7
IPC codes: G06G7