Adaptive-filter single-bit digital encoder and decoder with adaptation control circuit responsive to bitstream loading

Abstract: An adaptive single-bit encoder and decoder has its adaptive function determined by dynamically dividing the message frequency band into delta-sigma and delta modulation regimes of operation. In a practical embodiment this is accomplished by varying the corner frequency of a variable-frequency low-pass filter (14, 14′) in a leaky integrator so that below the corner frequency the operation is that of delta-sigma modulation and above the corner frequency the operation is that of delta modulation. An adaptation control circuit (18, 18′) removes the clock signal component from the encoded bit stream to provide an analog signal representative of bit stream information or loading for use in generating the control signal. The analog signal is peak rectified (22, 22′), smoothed (24, 24′), and (optionally) non-linearly processed (26, 26′) to provide the control signal.

Description: The invention relates generally to adaptive "single-bit" (delta and delta-sigma modulation) digital encoding and decoding systems for high-quality audio signals in which the message frequency band extends from very low audio frequencies, in the order of 20 to 50 Hz, to about 15 kHz. However, the invention is not limited to such applications. In particular, the invention relates to such systems in which the adaptive function, by means of an adaptive filter, variably divides the message frequency band into delta and delta-sigma modulation regimes of operation. A simple single-integration delta modulator is a type of single-bit digital encoder that encodes an audio signal as a series of ones and zeros in such a way that the average number of ones over a short period of time represents the instaneous slope of the audio signal. Each one-bit word of the bit stream tells the delta modulation decoder to take one step up or down to reconstruct the audio. The size of this step is a parameter of the design: small steps give small quantization errors but limit the maximum slope of the signal, and steps large enough to accommodate high-frequency high-level signals produce large quantization errors. An adaptive delta modulator changes the step size dynamically, attempting to provide an acceptable compromise between quantization-error level and high-frequency signal handling ability. In a delta-sigma modulator, which is also a single-bit digital device, the average number of ones over a short period of time represents the audio signal itself instead of its slope. Thus, the delta-sigma modulator, unlike the delta modulator, has an overload characteristic that is independent of frequency. Adaptive delta-sigma modulation systems are also well known. A common way to implement delta modulation encoders and decoders is to obtain the integration function by using a fixed-frequency low-pass filter having a corner (pole) frequency low in the message signal band (300 Hz in a prior art high-quality audio delta-modulation system). Such an arrangement is sometimes referred to as a "leaky integrator": below the corner (turnover) frequency of the filter (the "leak" frequency or "leak time constant"), the modulator acts as a delta-sigma modulator and, above the corner frequency, the modulator acts as a delta modulator. In adaptive systems, the gain of the fixed-frequency low-pass filter integrator is varied to achieve the adaptation. It appears that circuit designers have used leaky integrators for two principal reasons: a leaky integrator, unlike a pure integrator, does not require infinite gain at low frequencies; and a leaky integrator, unlike a pure integrator, quickly dissipates errors in the bit stream due to its relatively short time constant. The document JP-A-56-112121 discloses an adaptive single-bit digital decoder for decoding a delta-modulated digital signal into an analog signal. The document also discloses an encoder employing such decoder as a local decoder. In this prior art the gain of an integrating circuit is adaptively varied in response to a control signal representing the amplitude slope of the analog signal. This is achieved by changing the integrating constant of the integrating circuit. The integrating circuit is a low pass filter the time constant of which is decreased with an increase of the amplitude slope. The document ELEKTOR,March 1981, pages 3-56 to 3-68: "Elektor Hi-Fi Kompander", discloses a noise reduction compander system for a tape recorder, which employs a pre-emphasis network in the input path to the compressor and a de-emphasis network in the output path of the expander. The document PROC.IEE, vol. 117, no. 10, October 1970, pages 1915-1921, discloses a companded delta-sigma modulator in which, different from a normal uncompanded delta-sigma modulator, the encoded binary signal is not subtracted directly from the input analog signal but is modified by extracting the analog envelope of the encoded signal which is then multiplied by the encoded signal to obtain the feedback signal. For the extraction of the envelope a circuit is used which comprises a band pass filter, a full-wave rectifier, an envelope detector and a power-law element. It is an object of the present invention to provide a decoder and an encoder of the kind mentioned which have an improved signal-to-noise performance, particularly with respect to low-frequency noise in the presence of high-level high-frequency signals while retaining the advantage of the decreased sensitivity of leaky integrators to integration errors. This object is achieved with a decoder as claimed in claim 1 and an encoder as claimed in claim 9. Preferred embodiments of the invention are subjectmatter of the dependent claims. The invention recognizes that the division of the message signal band into delta modulation and delta-sigma modulation regimes of operation is desirable, but that further improvement in performance and simplicity of operation is achieved by dynamically varying the frequency at which the message band is divided into the two regimes of operation. This is accomplished by dynamically varying the corner (pole) frequency of the leaky integrator as the adaptive parameter of the system and by allowing, under certain signal conditions, the corner frequency of the leaky integrator to assume frequencies relatively high in the message signal band. In any adaptive single-bit digital encoding and decoding system, circuitry is required to determine the amount of adaptation needed at any given time. Many prior art adaptive delta modulators operate in the digital domain to determine the required adaptation, using various bit-counting algorithms and circuits to increase the step size when long strings of ones or zeros are encountered in the encoded digital audio bit stream. Other prior art delta-modulator adaptation-control circuits operate in the analog domain, typically employing techniques similar to those used in the control circuits of analog audio compressors and expanders, including the use of "speed-up" networks to minimize overload at the onset of transients. The control circuit of the present invention, although operating also in the analog domain, recognizes that the encoded digital bit stream carries audio information particularly well suited for adaptive control and that the audio information can be simply derived and processed as an analog signal for use as an adaptation control signal. In accordance with the invention, the adaptive function in a single-bit digital encoding and decoding system is provided by dynamically dividing the message frequency band into first and second regimes of operation in response to an adaptation control signal, the adaptation function in the portion of the message frequency band subject to the first regime of operation acting in the manner of delta-sigma modulation, and the adaptation function in the portion of the message frequency band subject to the second regime of operation acting in the manner of delta modulation. In a preferred embodiment, this is implemented by varying the low-pass filter corner frequency of a leaky integrator. Thus, the arrangement operates as a delta-sigma modulator below the filter corner frequency and a delta modulator above that frequency. The corner frequency of the variable low-pass filter, under certain signal conditions, assumes frequencies relatively high in the message frequency band in comparison to the low fixed corner frequencies (near the bottom of the message band) typically used in prior art leaky integrators. According to the present invention, a variable-frequency single-pole low-pass filter performs the adaptive function by adjusting the transition frequency between the delta and delta-sigma regimes in response to a control signal. There is no requirement to vary the gain of the integrator. Thus, the variable low-pass filter, or variable "leaky integrator," simply and inexpensively provides the delta-modulation integration function at frequencies above its variable cutoff frequency, thereby providing an integrator whose integration frequency changes but whose gain at frequencies below the filter cutoff frequency does not change. As the cutoff frequency shifts, the integration gain above the cutoff frequency changes due to the shifting of the single-pole low-pass filter characteristic, which rolls off at 6 dB/octave, in the preferred embodiment. This novel arrangement is believed to provide improved signal-to-noise performance, particularly with respect to low-frequency noise in the presence of high-level high-frequency signals, by taking advantage of the characteristics of delta and delta-sigma modulators and demodulators in response to dynamically changing signal conditions. The arrangement also retains the advantage of the decreased sensitivity of leaky integrators to integration errors. In addition, in practical circuit implementations, since variable-frequency low-pass filters have constant gain at low frequencies and DC, offsets at the input of the adaptive filter circuit are not likely to cause audible thumps when the filter frequency changes rapidly. In a preferred embodiment, the variable frequency low-pass filter is realized by placing broad-band negative feedback around an integrator whose gain can be varied. At frequencies where the forward gain through the integrator exceeds the feedback factor by a sufficient margin, the frequency response is the inverse of that of the feedback network--a flat response. At higher frequencies, the forward gain of the integrator is smaller than the feedback factor, and the overall response becomes that of the integrator. In this way the frequency at which the response changes from flat to integration changes as a function of the gain of the integrator: the circuit functions as a variable-frequency single-pole low-pass filter. In a preferred embodiment, the circuit includes an additional shelving low-pass filter in cascade with the variable filter. This additional filter serves both as a fixed pre- (in the encoder) and de-emphasis (in the decoder) network, reducing the audibility of noise modulation, and as a second integrator, providing noise shaping and improving the idle pattern of the modulator. In addition, in the preferred embodiment of the encoder, a further shelving low-pass filter is located in the feedback loop to the input comparator for the purpose of additional noise shaping. According to an embodiment of the invention, an adaptation control-signal generator measures the effective "bit-stream loading" to generate a control signal for adjusting the variable low-pass filter frequency. Bit-stream loading is an indicator of the amount of information carried in the encoded audio bit stream. The adaptation control-signal generator according to this invention operates in the analog domain to provide a simple, yet effective, means to maintain a high bit-stream loading. In addition, the adaptation control-signal generator of this aspect of the invention is inherently capable of quick response due to the nature of its rectifier and smoothing circuits, and does not need extra "speed-up" networks used in certain prior art adaptation circuits to minimize overload at the onset of transients. This aspect of the invention is also usable with conventional delta-modulation systems, including the aforementioned variable-gain, fixed-frequency leaky integrator arrangement. Because a delta-modulation signal can be decoded using an integrator, the audio content of a delta-modulation bit stream can be thought of as the time-derivative of the decoded audio signal produced when that bit stream is integrated. The derivative of an audio signal is, of course, its slope. This aspect of the invention is based on the recognition that because an adaptive delta modulator adjusts step size in response to changes in slope (an ideal adaptive delta modulator, for example, always accommodates the slope of an applied audio signal), the audio content of the bit stream, an indicator of slope, is well suited for use in adjusting the slope handling ability of a delta modulation system or a hybrid delta/delta-sigma modulation system. As the audio content of the bit stream (the bit-stream loading) rises, it indicates the presence of more and more information in the bit stream and that the delta modulator is approaching slope overloading. Ideally, the information in the bit stream is maximimized without slope overloading. In accordance with the teachings of this aspect of the invention, the audio content of the bit stream is transformed to the analog domain, rectified and smoothed. Optionally, the rectified and smoothed signal is then applied to a non-linear circuit such as an exponentiator, square-law, or cube-law device, which provides the control signal to the adapting element in a feedback-loop arrangement. In some applications, the non-linear circuit may be omitted. Either with or without the non-linear circuit, an analog audio compressor-type action is achieved. With the non-linear circuit, as the bit-stream loading increases to higher and higher levels, the degree of adaptation increases at an ever increasing rate and approaches a finite limit analogous to pure limiting in a compressor. In the case in which the non-linear circuit is omitted, a constant compression ratio results, determined by the gain in the control circuit. In a preferred embodiment of this aspect of the invention, the audio content of the bit stream can be transformed to the analog domain by applying the encoded bit stream to a fixed low-pass filter having a corner frequency just above the message frequency band, but below the clock frequency used to produce the bit stream, so that the clock signal component, at a frequency above the audio band, is removed. The filter output, an analog signal, is peak rectified, smoothed by a non-linear time constant and applied to the optional non-linear circuit for application to the control input of the variable low-pass filter. The filter cutoff frequency is a linear function of the applied control signal. As the peak bit-stream loading increases, the cutoff frequency of the variable low-pass filter increases. Brief Description of the Drawings Figure 1 is a functional block diagram of a preferred embodiment of the invention embodied in a modulator or encoder. Figure 2 is a functional block diagram of a preferred embodiment of the invention embodied in a demodulator or decoder. Figure 3 shows a set of theoretical response curves (gain versus frequency (logarithmic scale)) for the variable-frequency filters 14 and 14′ of Figures 1 and 2, respectively, showing, in solid lines, the quiescent low-frequency extreme position of the filters and the high-frequency extreme position of the filters, which is approached as a limit, and, in dashed lines, several examples of intermediate positions. Figure 4 shows the theoretical response curve for the single-pole low-pass shelving filters 16 and 16′ of Figures 1 and 2, respectively. Figure 5 shows the theoretical response curve for the single-pole low-pass shelving filter 17 of Figure 1. Figure 6 shows the theoretical combined response curve for the single-pole low-pass shelving filters 16 and 17 of Figure 1. Referring now to the drawings, Figure 1 shows a functional block diagram of a preferred embodiment of the invention embodied in a modulator or encoder, incorporating both the first aspect of the invention, the variable-frequency low-pass adaptive filter, and the second aspect of the invention, the bit-stream loading responsive adaptation control-signal generator. Figure 2 is a functional block diagram of a preferred embodiment of the invention embodied in a demodulator or decoder. Because the encoder includes the decoder, only the encoder of Figure 1 will be described in detail. Corresponding functional elements of the decoder that are also present in the encoder generally are labeled with the same reference numeral but with a prime mark ("′") in the decoder of Figure 2. Referring to Figure 1, the analog audio input is applied to input terminal 2 which is fed to a subtractor 6 that calculates the difference between the applied input audio signal and the reconstructed analog audio signal at the output of the encoder's local decoder. This difference is quantized to one of two binary levels by an operational-amplifier comparator 8 and is then time sampled by a D flip-flop 10 which is clocked by the clock input at terminal 4. The D flip-flop 10 provides the encoded digital signal output to output terminal 12 and to the variable-frequency low-pass adaptive filter 14. Subtractor 6, filter 17, comparator 8, and flip-flop 10 comprise digitizing means responsive to the two analog signals applied to subtractor 6 and to the clock sampling signal from terminal 4 applied to the flip-flop 10 for generating the digitally encoded output signal available at terminal 12; the output signal is a digital bit stream in which each single digital bit is responsive to a function (the filtered difference) of the two analog signals at each clock time. Figure 3 shows a set of theoretical response curves for the variable-frequency filters 14 and 14′, showing the low-frequency quiescent position and the high-frequency limiting position of the filter in solid lines and several examples of intermediate positions in dashed lines. The filter corner frequency, of course, may assume substantially an infinite number of positions in response to the DC control signal. The output of variable-frequency low-pass filter 14 is fed to a low-pass shelving filter 16 which provides the local analog decoder signal (available at terminal 11) for application to the inverting input of subtractor 6. Filter 16 provides a fixed pre-emphasis in the encoder; filter 16′ provides a fixed de-emphasis in the decoder. Both filters 16 and 16′ provide noise shaping and act as a second integrator. Figure 4 shows the theoretical response curve for the low-pass single-pole shelving filters 16 and 16′. The break frequencies f₁ and f₂ of the shelf response are discussed below. In the encoder of Figure 1, the filter 16 provides a pre-emphasis effect because it is contained in the feedback loop applied to subtractor 6. Thus, in the encoder its response provides a boost rising at 6 dB/octave between f₁ and f₂, which is complementary to the response shown in Figure 4. In the decoder of Figure 2, the filter 16′ provides a de-emphasis effect in accordance with the response shown, namely, a roll off falling at 6 dB/octave between f₁ and f₂. A further low-pass shelving filter 17 is provided in the encoder between subtractor 6 and comparator 8. Filter 17 provides additional noise shaping in the encoder, carrying the noise shaping down to a lower frequency than provided by filter 16. The placement of filter 17 is such that it has no effect on the pre-emphasis and has only a very small effect on the overload capability of the encoder. Figure 5 shows the theoretical response curve for the low-pass shelving network 17. The break frequencies f₃ and f₄ of the filter 17 shelf response are discussed below. Break frequency f₁ of filter 16 and break frequency f₄ of filter 17 are nearly the same. Thus, in the encoder, the cascade arrangement of filters 16 and 17 results in a combined response that is also a low-pass shelving characteristic. Figure 6 shows the theoretical combined response curve for filters 16 and 17. The digital bit stream from the D flip-flop 10 is also applied to the adaptation control signal generator 18 which applies a control signal to the variable-frequency filter 14 to control its corner (cutoff) frequency. The adaptation control signal generator 18 includes a low-pass filter 20, a full-wave peak-responding rectifier 22, a smoothing circuit 24, and (optionally) a non-linear circuit, a circuit embodying a function having a monotonically increasing slope, such as an exponentiating circuit 26. Although a square-law or cube-law circuit is suitable, in practice, an exponentiating circuit is used because such a circuit is easily implemented using bipolar transistors. Although improved performance is obtained when the optional non-linear circuit is used, in some applications it may be desirable to omit the non-linear circuit. In the decoder shown in Figure 2, the digital input 28 accepts a digital signal such as provided at output 12 of the encoder of Figure 1. The audio output signal at terminal 30 of the decoder is a close replica of the input signal at terminal 2 of the encoder of Figure 1 when the decoder receives an encoded digital signal from the encoder of Figure 1. A D flip-flop 10′ is provided in the input of the decoder of Figure 2 in order to provide a very clean digital bit stream to the adaptive filter 14′. The variable-frequency filter 14 may be implemented in various ways. One way is to implement it as a variable-gain integrator within a feedback loop. For example, a variable-gain integrator can be configured as an operational amplifier in combination with a variable transconductance amplifier operated as a variable current-gain amplifier. A suitable off-the-shelf variable amplifier is the RCA CA3280 integrated circuit chip which has two dual variable operational amplifiers. One of the two variable amplifiers may be used for the encoder while the other is used for the decoder. Such an arrangement is particularly useful when the encoder and decoder of the present invention are used, for example, in the same physical package to provide the digital signal for a digital delay line. In other applications, where the encoder and decoder are not on the same circuit board and do not share devices on the same integrated circuit chip, well-known component matching and temperature compensation techniques will likely be required. Also, the application and packaging of the practical implementation allows the use of a very high clock rate, in the order of 1 to 2 MHz. However, such a high clock rate is not critical to the invention. An external capacitor in parallel with the resistance of the internal linearizing diodes of the variable amplifier can be used to determine the pole of the low-pass shelving filter 16 (the break frequency f₁) and the combination of a resistor and another capacitor determine the zero (the break frequency f₂) of filter 16. It is recognized that the pole determined by the internal linearizing diodes of the variable amplifier and an external capacitor varies with signal level and temperature; however, in a practical implementation where the encoder and decoder are on the same printed circuit board and use two halves of the same integrated circuit device, there is very close tracking. In other applications, as mentioned above, additional component matching and temperature compensation may be necessary, using well-known techniques. With very high clock rates in the order of 1 to 2 MHz, the zero (the break frequency f₂ of filter 16) of the low-pass shelving network may be located above the audio spectrum, a zero frequency of about 20 to 30 kHz in a practical embodiment, allowing the full effect of the pre- and de-emphasis to be obtained. At lower clock rates, the zero must be moved down in frequency in order to keep the system stable. Although the exact break frequency is not critical, in a practical implementation of the preferred embodiment, 6 kHz was chosen as the pole frequency (the lower break frequency, corresponding to f₁ of filter 16 in Figure 1) of the pre and de-emphasis/noise shaping shelving network because it provides a reasonable compromise between adaptation range and signal handling capability. The rationale behind the choice of 6 kHz is based on the recognition that the spectral envelope of most music is such that, on the average, low-frequency components of music are at a higher level than high-frequency components and the components approximately follow a curve that rolls off at a 25 microsecond rate--a single-pole rolloff at 6 kHz. Thus, a pre-emphasis curve with a single pole at 6 kHz "pre-whitens" the music signal: it pushes up high-frequency components so that the resulting signal has approximately a white (flat) spectral balance. Although the pre-emphasis would appear to increase the probability of overload at high frequencies, this tends not to occur because most music does not contain high-level high-frequency components that would overload the system even after pre-emphasis. In a practical embodiment, the variable-frequency low-pass filter has a corner frequency that approaches 6 kHz as a theoretical limit. Thus, the overall effect of that variable filter in cascade with the pre-emphasis/de-emphasis shelving network is a 2-pole roll off at frequencies above 6 kHz. While the 2-pole rolloff provides a much better reduction in high-frequency noise modulation than does a single-pole roll off, it does adversely affect the high-frequency overload capability. However, this can be recovered by sacrificing some of the dynamic range, which is much more than adequate. One practical implementation of the invention has a dynamic range in excess of about 110 dB, thus allowing the high-frequency overload capability of a single-pole pre-emphasis to be regained by reducing the nominal input level to the circuit by the same amount as the additional boost at the top end of the audio spectrum (15 kHz); about 8 dB. The feedback path alters the mode of operation of the circuit, turning the variable-gain integrator into a variable-frequency integrator; without that feedback path the circuit would act as a conventional variable step-size delta modulator with additional noise shaping provided by the low-pass shelving network. The turnover frequency at which the variable-frequency integrator switches from a delta modulator to a delta-sigma modulator changes with the adaptation control signal. The effect of an external capacitor in the feedback path of the operational amplifier that operates in combination with the variable-gain amplifier in cooperation with feedback applied around the combination of the variable-gain amplifier and the operational amplifier provides the pole that forms the variable-frequency leaky integrator. Feedback applied around this network limits the low-frequency gain to a fixed value: at low frequencies when the loop gain of the feedback path around the combination of the operational amplifier and the variable-gain amplifier is larger than unity, the frequency response is flat as determined by a resistor in the feedback path. At high frequencies when the loop gain falls below unity, the overall gain is essentially the open-loop gain because the contribution of the feedback resistor becomes negligible. In that way, the result is a low-frequency pole that changes in frequency because the low-frequency gain remains constant and the amount of forward gain changes. The point at which the fixed gain caused by feedback intersects the changing open-loop gain causes the pole to move as the forward gain changes. In a practical implementation of this sort, the range of adaptation of the filter is limited such that the highest frequency that the filter can attain is approximately equal to the pole frequency of the shelving network. This provides a reasonable compromise between adaptation range and signal handling capability. If the pole frequency of the shelving network were lowered, then the maximum signal level curve would move down in frequency, thereby reducing the high-frequency signal handling capability relative to lower frequencies. The circuit operates so that, on a steady-state basis, the variable-frequency filter corner frequency substantially matches the frequency of an applied sine wave at a maximum amplitude level. If the applied sine wave has an amplitude level less than the maximum level for which the circuit parameters are chosen, the filter corner frequency does not fully shift to the frequency of the sine wave but is lower in frequency; the filter adopts a corner frequency lower than the sine-wave frequency in order to provide sufficient encoder gain at the sine-wave frequency to maintain proper bit-stream loading. A further advantage of this arrangement is the substantial elimination of audible "thumps" that occur in prior art variable step-size delta modulators, including not only those that use variable-gain integrators but also those in which pulse height is modulated. For example, in the arrangement of the present invention, the gain does not change at low frequencies and at DC; thus, there is no changing amplification of input offsets of a voltage-controlled element as there is in prior art variable-gain integrator type delta modulators. The circuit of the present invention may be built without any offset trims at the variable amplifier. As mentioned above, clock rates of about 1 to 2 MHz are suitable when the encoder and decoder are used in the environment of a digital audio delay line. However, much lower clock rates are usable for many applications and the clock rate is not critical to the invention. In an application such as a delay line where the encoder and decoder are in very close proximity, both the encoder and decoder flip-flops 10 and 10′ may be contained on a single integrated circuit chip. In many applications the low-pass filter 20 may be configured as a single-pole RC filter. However, in applications where a lower clock rate is used, it may be desirable to use second-order low-pass filters. The filter corner frequency may be located at or somewhat above the upper limit of the audio band handled by the overall system. For example, a main filter corner frequency of about 23 kHz is suitable for use in a system intended to handle signals as high as about 15 kHz. The low pass filter filters out the clock signal component to provide output a continuous analog signal representing the encoded audio in the digital bit stream. Because the flip-flop outputs are used as analog signals, noise filtering of the power source should be used. Rectifier 22 preferably charges the smoothing capacitor rapidly in response to peaks in the low-pass filtered bit stream. Smoothing circuit 24 has a fast attack/slow decay characteristic. In a practical implementation, about a 10-1 decay to attack time ratio is used. The adaptation control signal generator 18 preferably includes a means to set a threshold level above which the control signal adjusts the position of the variable-frequency filter 14 and below which level the control signal does not affect the variable-frequency filter. Above the threshold level, as the control signal (a current, for example) increases as an exponential function (as a result of block 26) of the filtered, rectified, and smoothed bit stream signal, the filter corner frequency shifts upward in frequency. Because the exponentiator is a non-linear circuit in series with the feedback loop of the control circuit, it causes more and more loop gain to be applied as the signal level rises. As the bit stream loading increases, the compression ratio increases so that at sufficiently high bit-stream loading levels it acts in the manner of an automatic gain control circuit with an infinite compression slope. At low levels, the compression slope is relaxed, reducing the chance of mistracking. The decoder also uses all of the circuitry described so far, but the encoder requires some additional circuitry: subtractor 6, which can be implemented as an operational amplifier, calculates the difference between the applied input audio signal and the reconstructed audio at the output of the encoder's local decoder. This difference is quantized to one of two binary levels by comparator 8, and is then time sampled by D flip-flop 10. The analog audio input to the encoder is applied at input 2 which couples the signal to an input of the operational amplifier functioning as the subtractor 6. The reconstructed audio signal from the filter 16 is applied to the operational amplifier and the polarity of the reconstructed audio signal with respect to the input audio signal results in subtraction. The noise-shaping characteristic of the low-pass shelving network 16 of Figure 1 can by improved by extending its low-frequency response. This is the function of filter 17, an RC network in the feedback path of the operational-amplifier subtractor 6. This provides a shelving network whose pole frequency (break frequency f₃ of filter 17 of Figure 1) equals that of the open-loop response of the operational amplifier, about 50 to 100 Hz, in a practical embodiment. A series resistor and capacitor in the operational amplifier's feedback path provide the zero (break frequency f₄ of filter 17) of the shelving network, about 6 kHz, in a practical embodiment. By locating the pole and zero creating elements at this point in the circuit rather than at the output (in association with variable filter 14), only the noise shaping characteristics and not the pre-emphasis are affected; thus the overload margin of the circuit is substantially unaffected. The discussion of the circuit operation thus far has considered mainly steady-state rather than dynamic operation. Because the arrangement of the present invention is an output-controlled feedback-loop system, it is susceptible to many of the same dynamic signal effects as audio signal compressors and expanders, for example, signal overshoots in compressors and signal undershoots in expanders. If an audio signal with a rapidly rising envelope is applied to the encoder of the present invention, the headroom provided above the steady-state adaption level determines how much level is available for the system to overshoot without overloading the bit stream. Adjusting the threshold in the adaptation-level control-signal path allows selection of the amount of headroom between the value of the steady-state adaptation level and the overload point of the bit stream. Ideally, the delta modulator as described operates to optimize the bit-stream loading. That is, the variable-frequency low-pass filter corner frequency is selected so as to maximize the information in the bit stream without overloading or clipping. When the bit stream overloads it puts out long streams of ones or zeros, indicating that it is slewing as far positive or negative as possible. Under these conditions, the low-pass filtered bit stream has the appearance of a clipped waveform, e.g. a straight horizontal line, either positive or negative. If such a clipped signal is applied, for example, to a fixed-gain integrator, the integrator output is a constantly rising or falling voltage--it is slew rate limited. By adjusting the attack time of the non-linear smoothing network in the adaptation level control signal circuit, the attack time of the system can be set so as to prevent the overshoot from exceeding a maximum magnitude. Adjustment of that attack time along with steady-state adaptation level allows the system to accommodate likely overshoots without overloading or clipping the bit stream excessively.

Claims: An adaptive single-bit digital decoder operating in a message frequency band for generating an analog output signal in response to a digital bit-stream representing an encoded input signal, said decoder comprising: control signal generating means (18; 18') responsive to the digital bit-stream for generating an adaptation control signal, adaptation means (14; 14') comprising variable-frequency low-pass filter means receiving said digital bit-stream, wherein the corner frequency of the low-pass filter means is varied in response to said adaptation control signal for dynamically dividing the message frequency band into first and second regimes of operation, the adaptation means in the portion of the message frequency band below the corner frequency and subject to said first regime of operation acting on said digital bit-stream in the manner of delta-sigma modulation, and the adaptation means in the portion of the message frequency band above the corner frequency and subject to said second regime of operation acting on said digital bit-stream in the manner of delta modulation,for generating an analog signal, wherein said adaptation means (14') is designed such that the portion of the message frequency band subject to said first regime extends, on a steady state basis, up to the frequency of an applied sine wave having an amplitude equal to a maximum amplitude level of the decoder, and output means (11; 30) for receiving the analog signal produced by the adaptation means and providing the analog output signal. A decoder according to claim 1 wherein said low-pass filter means is a single-pole filter. The decoder of claim 1 or 2, wherein a first noise shaping filter means (16; 16') for modifying the frequency response of the decoded analog signal is interposed between said low-pass filter means (14; 14') and the output means (11; 30), the first noise-shaping filter means having a low-pass shelving response with a pole frequency (f1) and a zero frequency (f2), the pole frequency (f1) being approximately the same as the maximum corner frequency of said low-pass filter means (14; 14'). The decoder of any one of the preceding claims, wherein the control signal generating means comprises means (20; 20') for removing a clock sampling signal component from the digitally encoded input signal to provide an analog signal representative of the amount of information carried by the digitally encoded input signal, means (22; 22') for rectifying the analog signal derived from the digital bit stream, and means (24; 24') for smoothing the rectified signal. The decoder of claim 4, wherein the means (22; 22') for rectifying the analog signal derived from the digitally encoded input signal comprises a peak-responding rectifier. The decoder of claim 4 or 5, wherein the means (24; 24') for smoothing the rectified signal includes a time-constant circuit having a fast charge and slow discharge time. The decoder of any of claims 4 to 6, wherein the control signal generating means further comprises means (26; 26') for non-linearly processing the smoothed signal. The decoder of claim 7, wherein the means for non-linearly processing the smoothed signal comprises an exponentiating circuit. An adaptive single-bit digital encoder operating in a message frequency band for generating a digitally encoded output signal representative of and in response to an analog input signal and a clock sampling signal, the encoder comprising digitizing means (6,8,10) responsive to the analog input signal, a reconstruction of the analog input signal, and to said clock sampling signal for generating the digitally-encoded output signal, the output signal being a digital bit-stream in which each single digital bit is responsive to a function of the analog input signal and the reconstruction of the analog input signal at each clock sampling signal sample time, and a local digital-to-analog decoder which receives the output signal as its digitally-encoded input signal and the analog output of which is used as the reconstruction of the analog input signal, characterized in that the decoder of any of claims 1 through 8 is used as the local digital-to-analog decoder. The encoder of claim 9, wherein the digitizing means includes further noise-shaping filter means (17), having a low-pass shelving response with a pole frequency (f3) and a zero frequency (f4), the zero frequency (f4) being approximately the same as the pole frequency (f1) of the first noise-shaping filter means (16) whereby the two noise-shaping filter means together generate a low-pass shelving response extending to a lower frequency than that of the first noise shaping filter means (16).

Inventors: MANDELL DOUGLAS EVAN, MANDELL, DOUGLAS EVAN
Applicants: DOLBY LAB LICENSING CORP, DOLBY LABORATORIES LICENSING CORPORATION
Application year: 1988
Main classification: H03M3
IPC codes: H03M3, H03H21, H03H17