TWO STAGE ADDRESS DECODER CIRCUIT FOR SEMICONDUCTOR MEMORIES

Abstract: The disclosed two stage address decoder circuit (AD) for 1/64 decoder operation includes a first stage comprised of two predecoder circuits (PDP; PDQ) operable to develop predecoded output signals (P0, ..., P7; Q0, ..., Q7) in response to input address signals (A1, ..., A3; A4, ..., A6) and the corresponding inverted address signals. Each predecoder circuit consists of a low power high speed Differential Cascode Current Switch (DCCS) tree with its associated current source. The second or final decode stage (FD) is comprised of a plurality of final decoding circuits (FD1, ..., FD64). Each FD circuit consists of a 2 way OR gate dynamically activated through a switched current source. The inputs of the 2 way OR gate are connected to one couple of said predecoded output signals (P1, Q1; ...; P8, Q8). Final decoder circuits supplies a final decoded output signals (X1, ..., X64) to drive the word lines (WL1, ..., WL64) of a memory cell array (MA). The switched current source is triggered by a control signal (SWL) supplied by a clock generator (CG) so that the FD circuits consume power only when the control signal is active. The use of a DCCS tree in the predecoder and of a switched OR gate in the final decoder circuit gives the address decoder circuit a competitive speed-power product with a safe and glitchless behavior.

Description: The present invention relates to semiconductor integrated circuits and more particularly to a two stage address decoder circuit for high speed large capacity Static Random Access Memories (SRAMs). In a semiconductor memory such as a SRAM, there is provided a word address decoder circuit which generates select/unselect signals for appropriately selecting a plurality of memory cells in the same row of a memory cell array (in other words, those cells which are connected to the same word line), while the remaining cells remain unselected. The address decoder circuit receives word address signals, in true and complement form, of e.g. n bits, and generate one select signal among 2 n combinations of the address signals. High speed and large capacity SRAMs play a key role in the development of modern high performance computer systems. To achieve this goal, BICMOS circuits which merge bipolar and CMOS technology, are highly desirable to provide both bipolar and CMOS devices with the performance/cost trade off. As far as address decoder circuits are concerned, the present tendency is to combine a high speed low power multistage bipolar address decoder circuit with a CMOS or BICMOS driver to drive the heavy capacitive loads on the word lines of the memory array, typically comprised of CMOS 6-device SRAM cells. It is well known that Current Mode Logic (CML) technology provides an extensive family of logic circuits, and in particular address decoder circuits with high speed operation at moderately low power levels. This is also true for its various derivatives or variants: e.g. the Differential Cascode Current Switch (DCCS) logic, Emitter Coupled Logic (ECL), etc. As far as DCCS logic is concerned, one may refer to the article: "Generation of multiple single-phase outputs from a single DCCS decode circuit" by M.E. Cohen et al, published in the IBM Technical Disclosure Bulletin, Vol. 26, N° 7A, December 1983, pp 3503-3504. A good example of such a multistage bipolar address decoder in CML technology is given in the article: "BICMOS circuit technology for a high speed SRAM", by T. Douseki et al published in the IEEE Journal of Solid State Circuits, vol. 23, No. 1, Feb. 88, pp 68-73. This article shows a way to design a bipolar address decoder circuit for a BICMOS memory. The decoder circuit may be understood as being composed of three circuit stages connected in cascade: a predecoding circuit of the ECL collector dotting type (CD), a level shifting circuit of the standard emitter follower type (EF), and a main decoding circuit so called a series gate circuit (SG) in CML technology. The CD circuit is used as the first layer of decoding. According to an original feature of the CD circuit, it is not necessary to generate explicitly the true and complement address signals, because the address receiver is included in the pre-decoder. However, this is not a definite advantage. True and complement address signals are generally available in a current switch environment. Alternatively, it is also possible to replace the complement address signals by a fixed reference voltage if supply and signal voltages are adequate. For each address signal, a receiver of the current switch type generates current information corresponding to the true and complement values. In others words, instead of using a voltage, a current provides the pre-decoding information. The collectors are all dotted together and a decoded address will correspond to no current flow in any of them. The main disadvantage or drawback of this solution is that a kind of current mirror must be done in the CD circuit and this can lead to a poor control on the down voltage corresponding in that case to a non-selection. A second drawback resides in the fact that the input capacitance of the CD circuit will be very high since transistors must be connected in parallel to provide the currents representative of a true/complement address, which increases the number of transistors used in the circuit implementation. The purpose of the EF circuit is to generate predecoded signals with a number of Vbe DC shifts (depending on the number of diodes) in order to drive the SG circuit that follows. In others words, it operates as a level converter. This construction has some inconveniences, because it does not provide an efficient solution in optimizing said SG circuit. The third decoding layer consists of the SG circuit. Because, the signals that drive the SG circuit have already been predecoded by the CD circuit, a differential tree, where more than 2 transistors are connected in each branch, can be used. If this construction reduces the number of layers in the tree, it has a significant drawback in that the common emitter nodes are more heavily loaded, which in turn, results in slowing down the current switching. In addition, it is clear that the up level depends of Vcc (minus a number of Vbe's), while the down level is referenced to ground. Finally, the saturation control of the SG circuit is poor, and becomes even more acute when the supply voltages are reduced. All these circuits are biased between a positive supply voltage (VCC = 0V) and a negative supply voltage (VEE = -5,2V), all use standard current sources connected to VEE. Seen as a whole, the disclosed three stage address decoder is power consuming in that there is no means provided to cut down the DC power in any stage of the decoder, causing a continuous consumption of power in the quiescent state, even if the circuits are not selected. Secondly, it has a limited flexibility and adaptability to different memory sizes and organizations because of the close interrelationships existing between the three circuits which in turn, makes it difficult to increase the number of the tree levels if so desired. Further, there is a risk for the output transistors of the SG circuit to saturate, so leading to slower switching. It is an object of the present invention to provide a high speed low power two stage address decoder with no DC power consumption in at least one of the two stages in the quiescent state, to limit overall circuit power consumption. It is another object of the present invention to provide a high speed low power two stage address decoder the architecture of which, has high flexibility and modularity to allow different configurations to be easily realizable irrespective the memory size and organisation are. It is still another object of the present invention to provide a high speed low power too stage address decoder wherein there is a means in the current source to avoid saturation of the output transistors in order to increase the speed. At last, it is also an object of the present invention to provide a high speed low power two stage address decoder circuit having a simplified circuit arrangement which requires a reduced number of devices. These and others objects of the present invention are accomplished by a two stage address decoder circuit as defined in claim 1 for decoding n input address signals and n inverted address signals into 2 n decoded output signals. Essentially the decoder circuit includes: a first stage comprised of at least two predecoder circuits operable to develop predecoded output signals in response to input address signals and the corresponding inverted address signals. Each predecoder circuit consists of a low power high speed Differential Cascode Current Switch (DCCS) tree with its associated current source. The address decoder circuit further includes a second or final decoding stage comprised of 2 n final decoder (FD) circuits. Each FD circuit consists of a 2-way OR gate dynamically activated through a switched current source. The inputs of the 2-way OR gate are connected to one couple of said predecoded output signals. Final decoder circuits supply final decoded output signals to drive the word lines of a memory cell array. The switched current source is triggered by a control signal supplied by a clock generator so that the FD circuits consume power only when the control signal is active. Further preferred features of this invention are set forth in the appended claims 2-11. The invention itself, however, as well as other objects and advantages thereof, may best be understood by reference to the following detailed description of an illustrated preferred embodiment, read in conjunction with the accompanying drawings in which: Fig. 1 is a block diagram showing the general architecture of the two stage address decoder circuit of the present invention to provide 1/64 decode signal to the word lines of a SRAM. Fig. 2 is a circuit diagram showing a typical embodiment of the two stage address decoder circuit including a DCCS tree with an improved current source in the predecoding stage and a plurality of switched OR gates in the final stage. Fig. 3 illustrates typical waveforms in the data path of the two stage address decoder circuit of the present invention. Fig. 1 is a block diagram showing the general architecture of the address decode scheme to provide 1/64 decode signal to a SRAM, according to the teachings of the present invention. The two stage address decoder circuit AD is comprised of two main functional blocks. First, a predecoder block referenced PD, comprising two predecoder circuits PDP and PDQ, is designed to operate as the first stage/section. Input address signals A1 to A3 (and inverted signals) are applied to the address lines of the predecoder PDP. Predecoded output signals P1 to P8 are available at the outputs of the PDP circuit. Similar construction applies to predecoder circuit PDQ respectively with input address signals A4 to A6 and predecoded output signals Q1 to Q8. Predecoder circuits PDQ and PDQ are identical and decode one out of 3 true/complement input address signals. As known, a three bit address signal can be decoded to provide eight different output signals. Two stage decoder circuit AD also includes a final decoder block FD to operate as the second stage and comprises 64 final decoder circuits referenced FD1 to FD64. They are adequately driven by the combinatorial set of predecoded output signals generated through the predecoding stages. For example, predecoded output signals P1 and Q1 are applied to FD1, while predecoded output signals P8 and Q8 are applied to FD64. The output of each FD circuit is connected to a word line driver which is designed differently depending on wether the SRAM technology is pure bipolar or BICMOS. Each of the final decoder e.g. FD1 supplies a decoded output signal, e.g. X1, to a word line driver circuit, e.g. WLD1. All the 64 word line drivers WLD1 to WLD64 form word line driver block WLD. Word line drivers, e.g. WLD1, provide word line output signals, e.g. W1 on the word lines, e.g. WL1. The latter are connected in a standard way to the memory cells (not represented) of a memory cell array MA. It is to be noted that according to this modular organization, a single circuit AD including several predecoder circuits and different word line driver blocks may service a plurality (K) of memory arrays (so called subarrays). According to a key feature of the present invention, the predecoders are built with a DCCS tree associated with an improved current source, and the final decoder circuits are each constituted of an OR gate loaded by a switched current source. Finally, to ensure adequate timing and power saving, final decoder circuits FD1 to FD64 operate under the control of a clock generator CG which provides control signals from a clock signal CLK. The use of this CG circuit is recommended for optimized operation of address decoder AD to save power consumption. Basically, this generator delivers a Set Word Line (SWL) control signal, when the main clock signal CLK is activated. However, it will not be described here in detail, since it is a mere single shot circuit. All these characteristics will be now detailed in conjunction with Fig. 2. Fig. 2 shows in details a typical″ embodiment of one predecoder circuit, e.g. PDP and one final decoder circuit, e.g. FD1 of the address decoder AD of the present invention. As explained previously, predecoder circuit PDP is used for decoding three binary valued input signals A1 to A3 to one of eight binary valued output signals P1 to P8. As shown in Fig. 2, the predecoder circuit PDP comprises one logical DCCS tree referenced STP, and one current source referenced CSP connected to the tree. The tree consists of a plurality of cells where each cell comprises a pair of transistors. The cells are arranged in levels corresponding to the number of address signals, and where each level is associated with a different one of the address signals. Level 1, comprising one cell (Q1, Q2), is associated with address signal A3. Level 2 is associated with address signal A2 and includes two cells (Q3, Q4, Q5, Q6). Level 3 is associated with address signal A1 and comprises four cells (Q7, Q8; Q9, Q10; Q11, Q12; Q13, Q14). Collectors of output transistors at level 3 are loaded by respective resistors R7, ..., R14 connected to a first supply voltage VH and of the same value. The address signals supplied to each cell are differential (true/complement) input address signals. Each cell comprises a pair of switching elements shown as transistors, whose emitters are coupled together and whose collectors provide output signals of the cell which are connected to the following level as input signals. The differential address signals are applied to the bases of the transistors of the cells. Each transistor functions as a switch to establish a current path from a current source in level 1 or a cell to another cell in the next level. The tree structure, therefore, has the capability of establishing one of eight unique current paths depending upon the binary value of the address signals say bits: A1, A2 and A3. Each unique current path corresponds to one of the eight possible permutations which can be established for three binary valued input signals. When selected, an output transistor is turned on and then the corresponding predecoded output signal is at a down level, then the 7 others output transistors are cut off and the corresponding predecoded output signals is at VH. The above arrangement is well known to those skilled in the art. Usually, the current source CSP is simply formed by a single transistor connected between the common emitter of Q1 and Q2 and a second supply voltage GND while its base is connected to a reference voltage VREF. To improve speed, it is essential to keep output transistors at level 3, far from any risk of saturation. In that respect, the standard current source has been significantly improved. The usual single transistor is split into four elementary transistors T1.1 to T1.4 the base electrodes of which are tied together and connected to a reference voltage VREF supplied by a special antisaturation circuit operating in a current mirror mode. This unique antisaturation circuit includes transistor T0 mounted in a diode configuration since its base-collector junction is shorted, and loaded by resistor R0 connected to VH. Another load resistor RD in series with a diode D are connected in parallel with resistor R0. As a result, the STP tree shown in Fig. 2, is basically a 3 level DCCS circuit for pre-decoding the 3 address signals applied to the address lines. For that purpose the address signals are true and complement signals generated on m = 3 different referenced levels. As an example, these complement (or inverted) address signals can be generated through a standard differential amplifier (not shown), comprised of two bipolar transistors having an emitter coupled configuration. The collectors of these transistors are connected through load resistors to a supply voltage Vref(m). The address signal is applied to the base electrode of one transistor, while the other base electrode is connected to a reference voltage Vref. The characteristic voltage swing of the Ai and Ai (Address i true and Address i complement with i = 1 to m) is not greater than 300 mV. This very small differential signal helps to achieve fast operation with a confortable margin in noise immunity. It is well known that differential ECL type of circuit can be operated with small swing input signals whilst maintaining a sufficient DC noise tolerance. These 3 pairs of address signals A1 to Am (m = 1 to 3) are″ referenced with respect to GND at 3 different values. The value at which these signals are referenced is determined by starting to consider address signals A3/ A 3 . They must be as low as possible to allow later the largest swing at the collectors of the output transistors, keeping in mind that the current source transistors (T1.1 to T1.4) have to operate sufficiently far from saturation to prevent speed failure and down level output voltage degradation. Vref(3) has been chosen equal to 1.2 V. Then the up level on A3/ A 3 is 1.2 V and the down level is (1.2 V - 0.3 V = 0.9V), since 300 mV is our standard for Ai/ Ai voltage swing. This value of 1.2 V for Vref(3) provides a voltage of 0.4 V across the transistors T1.1-T1.4, which is sufficient to ensure a correct current mirroring of transistor T0. Each of the other reference voltages Vref(2), Vref(1) is successively determined with such a constant voltage increment. The latter must be as low as possible, to maximize the swing capability, but not too small to cause saturation of transistors, which in turn, could result in consequent speed degradation and failure in operation. For these reasons, the incremental voltage value between the different address signals in the DCCS tree has been chosen to be around 300 mv. The Vcb of any transistor in the DCCS tree is given by Vref(k)-Vref(k-1)-Vbe, with k = 1 to m. The 300 mV of voltage incremental between the Vref(m) allows an automatic clamp of about 0.5 V across the collector-base junction of any transistor in the tree, which keeps all transistors far away from saturation. With semiconductor technologies known to date and the continuous trend towards reduced power supplies (e.g. VH = 3.5V), the level number m, which also corresponds to the number of address signals to be decoded, such that m ≦ 5. High value resistors R (in the range of 10 k Ω) are provided at all the intermediate common emitter nodes in the DCCS tree to prevent these nodes to be charged to high value voltage through different leakage paths. Otherwise, this would lead to performance unpredictability, especially during the long periods of time the SRAM is not addressed. The design of the current source can be made differently depending on the type of cell that is used in the memory cell array. If the technology considered is a BICMOS one, the mirrored current in the transistor T0 is such that the predecoded output signals on the 8 collectors of the output transistors at the end of the DCCS tree will be driven with respect to GND at the down level. The respective high voltage on said signals will always follow VH. The goal of current source CSP is to drive the collectors of the output transistors (Q7 to Q14) as low as possible in order to be able to drive a BICMOS driver in the best conditions. It is well known that a BICMOS or a CMOS driver needs a large voltage swing to operate efficiently (overdrive Vgs above the threshold voltage). According to a significant feature of the present invention, it is possible to adjust the various resistors: RD, R0 and R7-R14, in order to get the desired down voltage at the collectors of the output transistors of the DCCS tree, this voltage being referenced to GND. If we assume that R0= k x RD and RP (P = 7 to 14) = k′RD , we can compute a given predecoded output signal, e.g. P1: P1 = VH-k′RD [( VH - Vbe ( Q 7))/ kRD + ( VH - Vbe ( Q 7) - Vbe ( Q 8))/ RD ] P1 = VH-k′RD[( VH (1 + k ) - Vbe (1 + 2 k ))/ kRD ] If we want to have P1 independent from VH variations and to adjust P1 at the down level to 1.7 Vbe; we must have: k/k′(1 + k) = 1 and 1.7 Vbe = k′(1 + 2k) Vbe / k Solving this system of equations, we find k = 1.5 and k′ 0.64. It is therefore possible to adjust the parameters k and k′ to have 1.7 Vbe at the collectors of the output transistors of the DCCS tree (in that case k = 1.5, k′ =0.64). RD can be seen as a normalized value which determines the speed of the predecoder circuit. As a result, it is very easy to increase or to slow down the speed of the DCCS tree, since its performance is now mainly dependant on the value of RD. Still considering Fig. 2, any FD circuit, e.g. FD1, is basically a switchable OR gate built with emitter dot followers. The OR gate includes transistors T and T′ connected in parallel between VH and a common emitter or OUT node at the same potential as the OUT terminal. Said OUT node is connected to a switchable current source comprising transistor T1 and resistor R1 connected to GND. Signal SWL is applied at the base of transistor T1. Depending on the type of the world line driver circuit used, a single control signal SWL may be sufficient to activate the FD circuits, for example, if they are of the self power-off type. With standard word line drivers, a second control signal RWL (Reset Word Line) is necessary. Control signal RWL is also generated by the CG circuit. In this case, a third transistor T2 is mounted in parallel with transistors T and T′, its base being driven by said control signal RWL. Increased flexibility can be given to the OR gate if so desired. For example, an additional transistor T″ (not shown) may be added in parallel with transistors T and T′ in the OR gate, if a supplemental predecoder/current source PDR/CSR (not shown) is desired to increase the number of decoded output signal/lines. Therefore, SWL and RWL control signals are respectively used, to set a decoded word line within the cell array during a predetermined time window, and to turn off the 64 FD circuits. The word line that is selected corresponds to the set of collector outputs that are low. The address is kept stable during the READ operation, the end of which is conditioned by the RWL control signal. This will become even more apparent with the description of Fig. 3. Fig. 3 shows typical waveforms in the data path of the two stage address decoder circuit of the present invention. Curves 31a and 31b show the address input signals An and An and illustrate the voltage difference of 0.3 V used to distinguish the true/complement signals. Curve 32 shows one typical predecoded output signal P1 to P8 or Q1 to Q8. Curve 33 is the SWL control signal. It is a calibrated pulse related to the total memory cycle; its width determines the power dissipation of the final decoder circuits. Curve 34 shows a typical decoded output signal X1 to X64. Curve 35 is the RWL control signal. It is also a calibrated pulse, but its width is not critical to the overall circuit operation. In fact, only the rising edge of the signal is used to restore the output signals to the high voltage: VH-Vbe (T2). A typical word line output signal W1 to W64 is illustrated by curve 36. TW represents the duration of the memory array selection window. This window is determined by the rising transitions of SWL and RWL. It is important to notice that the minimum SWL pulse width has to be adjusted in order to have a correct voltage position of decoded output signals X1 to X64. In summary, the operation of address decoder AD is as follows. The address signals propagate through the DCCS tree and finally only 1 out of 8 collector outputs in the DCCS tree referenced STP will be at a down level since in our case, a deselection in the DCCS tree corresponds to a high voltage value which is given by the power supply VH. The clock generator CG generates the SWL control signal, so that a current can flow in transistors T1.1 to T1.4. The decoded output signal of the one FD circuit that has a low voltage on each base of the transistors T1.1 to T1.4 will discharge one Vbe below the down voltage provided at the output of the DCCS. Control signal SWL can only be activated for a time period compatible with the total capacitance connected at the OUT terminal. By this way it is possible to minimize the power consumption in the final decoder stage. Next, the RWL control signal is applied at the base of transistor T2 and restores the OUT node to VH-Vbe(T2). As explained above, K represents the number of partial cell arrays that can be driven (eventually in parallel). There is no doubt that the value of K will impact the performances at a given power consumption, but the DCCS pre-decoder will continue to operate normally under a wide range of K. It is also obvious that the clock signals must be optimized differently depending on K. When K increases, the SWL and RWL control signals must be″ delayed. In a more generic sense, the decoding scheme of the present invention may be extended to the use of several predecoder circuits: PDP, PDQ, PDR, ... to decode m, m′, m″, ... address signals so that n = m + m' + m'' . The respective number of predecoded output signals is therefore M = 2 m , M' = 2 m′ , M'' = 2 m'' and the number of final decoders is given by N = 2 m x 2 m′ x 2 m'' x ... or N = 2 m + m′ + m″ . For example, n=6, m=m′=3, we obtain N = 2 m + m′ = 2 n = 64 which corresponds to the implementation shown in Fig.2. With n=10, m=4, m′=4 and m''=2, we obtain N=1024. This clearly demonstrates that the present decoding scheme is very powerful and flexible. The advantages of the two stage address decoder of the present invention may be summarized as follows: Low power/high speed DCCS predecoder, wherein the performance is almost only dependent on the normalized value of resistor RD. The signal swing amplification is made in the first stage circuit so that there is no need for a specific signal conversion in front of the word line driver. No DC power is consumed in the second stage, because the FD circuit is of the switched type. In addition, it provides a mean to suppress any address glitch problems. This advantage results from combining an OR gate with a switched current source activated by a control signal SWL of the pulse type, so that power consumption minimized. Different configurations are easily realizable when the number of predecoder circuits and/or the number of levels increases. According to the DCCS tree implementation, it is possible to drive either an ECL or a BiCMOS word line driver. In the field of BiCMOS application the DCCS tree provides a signal which is referenced to VH and GND, while in the field of ECL application, the DCCS tree output signal can be referenced with respect to VH only. Voltage Vref(m) allows the adjustment of the operating point in the DCCS over a wide range. Although the above description was made with reference to an address decoder circuit comprised of two 3 level DCCS predecoder circuits with sixty-four FD circuits, it is to be″ understood that the present invention is not limited to, but has a wider range of applications according in particular to the speed/power target that must be reached for the whole memory. This invention is widely applicable, not only to a″ semiconductor memory such as a dynamic RAM, a static RAM or any of various ROMs, but also to various semiconductor integrated circuits each comprising a decoder circuit which receives address signals composed of a plurality of (n) bits and forms 2 n predecoded output signals. Although the invention was described with reference to a word address decoder, it is also applicable to a column address decoder.

Claims: A two stage address decoder circuit (AD) for decoding n input address signals (A1... An) and n inverted address signals into 2 n decoded output signals, comprising: a first stage biased between a first supply voltage (VH) and a second supply voltage (GND), including at least two predecoder circuits (PDP, PDQ, ...) each consisting of a multilevel Differential Cascode Current Switch tree (STP, ...) and a respective constant current source (CSP, ...) for decoding respectively, a separated group (A1, ... Am; Am + 1, ... Am + m', ... An) of m said input address signals and generating a respective group of 2 m predecoded output signals (P1, ...., P(2 m ); Q1, ..., Q(2 m ' ); ...), where m is less than n; a second stage including 2 n final decoder circuits (FD1, ... FDN) driven by at least one of said predecoded output signals (P1, Q1 ...; P(2 m ), Q(2 m ' ), ...) from each group to provide 2 n final decoded output signals (X1, ..., XN). The two stage decoder of claim 1 further comprising a clock generating circuit (CG) to generate at least one first control signal (SWL) to dynamically activate said final decoder circuits. The two stage decoder of claim 2 wherein each of said final decoder circuits consists of an OR gate activated by a switched circuit which is triggered under the control of said first control signal (SWL). The two stage decoder of claim 3 wherein each of said final decoder circuits consists of first and second bipolar transistors (T, T') connected in parallel, each of the bases is driven by one of said predecoded output signals, the collectors of which are connected to the first supply voltage (VH) and the emitters of which are coupled together in a common node (OUT) which is connected to a switched current source emitter follower circuit. The two stage decoder of claim 4 wherein said emitter follower circuit comprises a third bipolar transistor (T1) connected in series with a resistor (R1) tied to the second supply voltage (GND), the base of the third transistor receives said first control signal (SWL) and its collector is connected to said common node. The two stage decoder of claim 5 wherein a fourth transistor is connected in parallel with said first and second bipolar transistors, the base of the fourth transistor is driven by a second control signal (RWL). The two stage decoder of any of the above claims wherein said Differential Cascode Current Switch tree is comprised of a plurality of cells where each cell comprises a pair of transistors with coupled emitters, and the collectors of output transistors (Q7, ..., Q 14) providing said predecoded output signals are loaded by respective resistors (R7, ..., R14) connected to said first supply voltage (VH). The two stage decoder according to claim 7 wherein each of said current sources (CSP, ...) is comprised of a plurality of bipolar transistors (T1.1 - T1.4) connected in parallel with their common emitter electrodes connected to said second supply voltage (GND), and their common base electrodes connected to a voltage reference generator consisting of a current mirror circuit. The two stage decoder of claim 8 wherein said current mirror circuit consists of a transistor (T0) mounted in a diode configuration and loaded in a first branch by a first resistor (R0) and in a second branch by a second resistor (RD) connected in series with a diode (D). The two stage decoder of claim 9 wherein the load resistors (R7-R14) connected to said output transistors in said tree and said first resistor are so arranged that the potential at the collectors of output transistors is clamped to 1,7 Vbe. The two stage decoder of any of the claims 7-10 wherein each pair of coupled emitters of each cell of said tree, other than the pair coupled to the constant current source, is connected to the second supply voltage (GND) through a resistor (R).

Inventors: LEFORESTIER, SYLVAIN, OMET, DOMINIQUE
Applicants: INTERNATIONAL BUSINESS MACHINES CORPORATION
Application year: 1988
Main classification: G11C8
IPC codes: G11C11, G11C8