Improved differential single-ended converter

Abstract: A differential to a single-ended converter circuit includes linearizing PN junctions that compensate for input signal voltage loss in the input stage. A variable voltage source and a resistor are provided for adjusting the voltage produced across these junctions by varying the current through these junctions. The converter circuit also includes a shunt feedback amplifier for inverting half the differential output voltage so that it sums with the differential voltage of opposite phase to produce the single-ended output signal. The shunt feedback amplifier is operated at a constant current to minimize error voltages. Bootstrapping is employed to linearize thermal distortion of the PN junctions and means are provided for compensating for the resulting thermal distortion.

Description: TECHNICAL FIELD This invention relates generally to high speed differential to single-ended converter circuits and more particularly to means for improving the linearity and minimizing the power consumption of such circuits. BACKGROUND OF THE INVENTION Differential to single-ended converter circuits receive a differential signal composed of two input signals and convert the input signals to a single-ended output signal. Such circuits find application in the output stages of analog integrated circuits wherever a differential signal must be converted to a single-ended signal. For example, a differential amplifier that provides signals to an analog-to-digital converter would include a converter circuit as an output stage because analog-to-digital converters are typically designed to receive only single-ended signals. Prior high speed converter circuits have a number of drawbacks. For one, they fail to maintain a linear relationship of output to input signal when the output signal voltage is measured relative to ground. For another, they tend to consume an undesirable amount of power to achieve linearity. Nonlinearity develops because of signal loss across PN junctions in the input stage and because of thermal distortion caused by the heating and cooling of such junctions during circuit operation. The undesirable power consumption arises from the increased standing current through the PN junctions required to improve linearity and from the increased gain needed in converting the signal. Typically, the output signal in conventional converters is measured with respect to ground and is proportional to only one of the input signals. In effect, half of the differential output signal is discarded. This loss in signal is compensated for by increasing the overall gain of the converter circuit, but at the cost of the increased power consumption. The additional power dissipated in turn causes more heating of the integrated circuit and increases the thermal distortion. An object of the invention, therefore, is to provide a differential to single-ended converter circuit that compensates for the nonlinear signal losser and for thermal distorsion. EP-A1-0127832 discloses a circuit for converting a differential input signal to a single-ended output signal. The circuit has first and second transistors with emitter electrodes connected together and to a constant current source through respective resistors. The input signal is applied to their base electrodes, and their collector electrodes supply respective collector currents. A dual-single converter circuit receives the collector currents at its input and supplies an output current. At least one variable current source is adjustable to vary the conversion factor of the circuit. US-A-4605906 discloses a differential amplifier circuit, although one which does not have a single-ended output. Linearity correction is provided by using compensation diodes and transistors in the respective load circuits of a pair of transistors the bases of which are connected to receive the differential input signal. In accordance with the present invention there is provided a circuit for converting a differential input signal to a single-ended output signal, the circuit including an input stage having PN junctions for receiving the differential input signal to produce a differential output current and having an output stage to provide the single-ended output signal, being characterised by: means coupled to the output terminals of the input stage for isolating the input stage to produce a buffered differential output current so that the voltage at the output terminals of the isolating means is isolated from the voltage at the input terminals of said means; PN junction means matched to the PN junctions of the input stage and coupled to receive the buffered differential output current for applying to the output terminals of the isolating means a compensated differential voltage equal to the sum of a portion of the differential input signal lost across the PN junctions, and of a differential voltage which is proportional to the buffered differential output current, the input of the output stage being coupled to one of the two output terminals of the isolating means; and means coupled to the output terminals of the input stage for adjusting the compensated voltage by adjusting the differential current received at the input terminals of the isolating means so that the current through the PN junction means is adjusted. In the preferred embodiment, adjustment of the compensating voltage enables the gain or transfer function of the converter circuit to be adjusted to be compressive, linear, or expansive. The converter circuit also preferably includes means for bootstrapping the linearizing PN junctions so that thermal distortion generated by self-heating of junctions varies linearly with the input signal. A thermal compensation means responds to the input signal to generate thermal distortion that compensates for the thermal distortion of the PN junctions. The compensation means may be used to provide compensation for other sources of thermal distortion within the circuit as well. The converter circuit includes amplifier means for converting the differential input voltages to differential output currents which are coupled through common base stages. One common base stage is coupled to an inverting shunt feedback amplifier to invert half the output voltage so that it sums with the output voltage of opposite phase. The difference portion of the input signals is therefore added to produce the output signal and the common portion of the input signals is therefore rejected from the output signal. Arrangements embodying the present invention will now be described by way of example with reference to the accompanying drawings. BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a schematic diagram of a differential to single-ended converter in accordance with the present invention. FIG. 2 is a graph of voltage gain as a function of differential input signal voltage. FIG. 3 is a schematic diagram of a differential to single-ended circuit similar to that in FIG. 1 with additional components added for higher gain and improved linearity. DETAILED DESCRIPTION Referring now to FIG. 1 of the drawings, there is shown a differential to single-ended converter circuit according to the invention. The circuit comprises a number of stages for converting a differential signal composed of two opposite phase input signals into a single-ended output signal. Through these stages the output signal, whose voltage is measured with respect to ground, is linearly related to the difference between the input signals. A differential input stage of the converter circuit comprises a pair of opposed transistors Q₁ and Q₂ whose emitters are coupled together through a resistor R₁. The emitters of transistors Q₁ and Q₂ are also each coupled to a current source stage comprising transistors Q₁₀, Q₁₁, current setting resistors R₆ and R₇, and a negative voltage source V EE . Transistors Q₁₀, Q₁₁ provide the necessary standing or quiescent current I M for each of the transistors Q₁ and Q₂, respectively. The base of transistor Q₁ is connected to an input 12 of the input stage to which an input signal voltage +V IN is applied; similarly, the base of transistor Q₂ is connected to an input terminal 14 to which an input signal -V IN is applied. Thus, +V IN and -V IN constitute the differential input signal, herein a voltage. The differential signal voltage is applied across the base-emitter junctions of transistors Q₁, Q₂ and across resistor R₁, producing differential output currents +I DO and -I DO . The total current in each collector path of transistors Q₁ and Q₂ thus equals the standing current I M and the differential currents +I DO or -I DO as indicated in FIG. 1. Coupled to the collectors of transistors Q₁ and Q₂ is a means for isolating the differential input stage from the output signal. In the present embodiment, such means comprises a conventional common base stage having wide-band near unity current gain. The common base stage includes a pair of opposed transistors Q₃ and Q₄ to whose respective bases is applied a constant base voltage V B . The emitters of transistors Q₃ and Q₄ are coupled to the collectors of transistors Q₁ and Q₂ respectively to pass the differential output currents +I DO and -I DO to further stages in the converter circuit. The transistor Q₄ isolates at its collector the voltage swing in the single-ended output signal from the voltage signal at the collector of transistor Q₂. This isolation minimizes the Miller effect that otherwise would develop across the collector-base junction of transistor Q₂. A balance is thereby maintained in the input signal voltages at the collectors of transistors Q₁ and Q₂. As indicated, the differential output currents +I DO and -I DO sum to the differential current produced by the application of the input signal voltage across the base-emitter junctions of transistors Q₁, Q₂ and resistor R₁. Because of the nature of the differential signal and the input stage, differential currents +I DO and -I DO are equal but opposite in sign. These differential currents are passed from transistors Q₃, Q₄ through resistors R₂ and R₃. The resistors R₂ and R₃ alone, however, do not linearly reproduce the input signal because of the voltage lost across the base-emitter junctions of transistors Q₁ and Q₂. To compensate for that loss, PN junction means such as transistors Q₅ and Q₆ are incorporated into the circuit, matched to transistors Q₁ and Q₂, respectively. The emitters of transistors Q₅ and Q₆ are serially connected to the resistors R₂ and R₃, respectively, to reproduce the voltage across the base-emitter junctions of transistors Q₁ and Q₂. The differential currents +I DO and -I DO now flow through the base-emitter junctions of transistors Q₅ and Q₆ as well as the resistors R₂ and R₃. The junctions produce a compensating signal voltage substantially equal to the signal voltage lost across the base-emitter junctions of transistors Q₁ and Q₂. With the addition of transistors Q₅ and Q₆, the output signal voltage is now a linear function of the differential input signal voltage. In FIG. 1, resistors R₂ and R₃ are each equal to one-half the resistance of resistor R₁ and therefore provide together with the base-emitter junctions of transistors Q₅ and Q₆ unity voltage gain from input to output. As will be seen in Fig. 3, other multiples of gain can be produced as well by increasing the resistance of resistors R₂ and R₃ and by adding additional PN junctions for linearization. FIG. 2 is a graph of voltage gain (A v = V OUT /V IN ) as a function of the input signal voltage swing across the base-emitter junctions of transistors Q₅ or Q₆. The junction is plotted from a -V, where transistor Q₁ shuts off, to a +V, where transistor Q₂ shuts off. The innermost curve 16 shows the voltage gain A v without the presence of the base-emitter junctions of transistors Q₅ and Q₆. The gain or transfer function is compressive (less than unity) until the input signal voltage is relatively close to the DC operating point. With transistors Q₅ and Q₆ present in the converter circuit, the gain is linear over a greater range as indicated by the middle curve 18. The voltage gain may also be made expansive (greater than unity) as indicated by line 20 to compensate for compression in circuits preceding the converter circuit. Means for so adjusting the compensating voltage produced across the transistors Q₅ and Q₆ comprise in FIG. 1 a variable voltage source V LIN for applying voltage cross resistors R₉ and R₁₀ to produce currents I LIN . These currents are directed into the emitter-collector couplings of transistors Q₁, Q₃ and Q₂, Q₄, respectively. The collector currents I DO and I M of transistors Q₁ and Q₂ are fixed by the input signal voltage and the current source transistors Q₁₀, Q₁₁. The currents produced by V LIN thus affect the amount of current drawn through transistors Q₅ and Q₆ into the collectors of transistors Q₃ and Q₄. As the current I LIN is increased, the differential currents through the base-emitter junctions of transistors Q₅ and Q₆ correspondingly decrease. Similarly, as the current I LIN is drawn out of the emitters of transistors Q₃ and Q₄ by changing the voltage applied across resistors R₉ and R₁₀, the differential output currents drawn through these base-emitter junctions correspondingly increase. These adjusting means provide the circuit designer with the ability to fine tune the linearity of the converter circuit. The voltages developed across resistor R₂ and the base-emitter junction of transistor Q₅ by +I DO must be inverted to add to rather than subtract from the voltage developed across resistor R₃ and the base-emitter junction of transistor Q₆ by -I DO . This inversion is accomplished through the shunt feedback amplifier that includes transistor Q₉, whose emitter is connected to ground and whose base is connected between the collector of transistor Q₃ and the resistor R₂. The collector of transistor Q₉ draws current through a current limiting resistor R₅, which also provides base current to transistors Q₅ and Q₆. Transistor Q₉ with its base-emitter junction tied to a fixed voltage such as ground maintains the voltage at the collector of transistor Q₃ constant, forcing the voltage at the emitter of transistor Q₅ to vary linearly in response to a change in +I DO . This can be seen by following the feedback loop through transistor Q₉. If V IN swings positive, for example, increased differential current +I DO is drawn through the collector of transistor Q₁, forcing the voltage on the collector of transistor Q₃ to decrease momentarily. This decrease in turn decreases the base-emitter junction voltage of transistor Q₉ and the amount of current drawn through the collector of transistor Q₉. The voltage on the collector of transistor Q₉ correspondingly rises as the collector current decreases, raising the voltage on the base of transistors Q₅ and Q₆. This rise in the base voltage raises the emitter voltage until the voltage across resistor R₂ increases to accommodate the increase in current +I DO . With the voltage across resistor R₂ and transistor Q₅ so inverted, the signal voltage measured across resistor R₂, transistors Q₅, Q₆, and resistor R₃ sums to substantially the input signal voltage between the input terminals 12 and 14. For example, with +V IN swinging positively and -V IN swinging negatively, the voltage at the collector of transistor Q₄ increases as -I DO decreases. The voltage on the base of transistor Q₆ rises because of the inversion at the connected base of transistor Q₅. This rise in the base voltage raises the emitter voltage. This in turn raises the voltage across resistor R₃, thus effectively adding the voltages. The result of this voltage addition appears at the collector of transistor Q₄ as the output voltage and causes the collector to swing from its DC operating point by that amount. The output voltage change is applied to the base of an emitter follower transistor Q₁₂ which provides a low output impedance for the circuit. The inversion of the voltage signal on the base of transistor Q₅ also provides excellent common mode rejection for the converter circuit. If +V IN and -V IN both rise, for example, the voltage at the base of transistor Q₅ will increase while the voltage across the base-emitter junction of transistor Q₆ and resistor R₃ will decrease a like amount. The net result is rejection of the common mode signal and no change in the output signal voltage at the collector of transistor Q₄. The transistor Q₉ also maintains the quiescent output voltage or DC operating point of transistor Q₁₂ at a predetermined voltage level, with transistor Q₁₂ preferably matched to transistor Q₉. This operating point is typically chosen to make the converter circuit compatible with other integrated circuits that may connect to the output of transistor Q₁₂. In the present embodiment, the DC operating voltage at the base of transistor Q₉ is one base-emitter junction above ground. This voltage is raised a second base-emitter junction through transistor Q₅ and then decreased two base-emitter junctions through transistors Q₆ and Q₁₂, respectively. This voltage path therefore places the DC output voltage of transistor Q₁₂ at ground. It should be understood, however, that the emitter of transistor Q₁₂ could be set within a range of DC operating voltages by changing the level of voltage that is applied at the emitter of transistor Q₉. To improve further the linear response of the converter circuit, the transistor Q₉ should be operated at a constant current to eliminate error voltages caused by bias changes of the base-emitter junction of transistor Q₉. A source of constant current is provided by the resistor R₅ that is connected at one end to the collector of transistor Q₉. With resistor R₄ equal to resistors R₂ and R₃, the voltage across resistor R₅ is kept nearly constant by tying the collector of transistor Q₆ to the junction of the emitter of a transistor Q₇ and resistor R₅. Now, the collector voltage of transistor Q₉ and the base voltage of transistor Q₅ rise by a voltage equal to: ΔV=ΔI DO (R₂)+ΔV BE of transistor Q₅ in response to an increase in +I DO . Current from the collector of transistor Q₆ (-I DO ) decreases a like amount in response to the input signal, causing the voltage across transistor Q₇ and resistor R₄ to decrease by: -ΔV=-ΔI DO (R₄)+ΔV BE of transistor Q₇ Therefore, the voltage applied across resistor R₅ is kept nearly constant because resistor R₄ equals resistor R₂ and -I DO is equal but opposite to +I DO . The current provided by the resistor R₅ to transistor Q₉ is therefore constant. Thermal distortion from the self-heating of the converter circuit also affects voltages and currents therein. Distortion due to simultaneous changes in voltage and current tends to be nonlinear; distortion due to only a voltage on a current change, however, tends to be linear. In the converter circuit, transistors Q₉ and Q₁₂ produce linear distortion because both operate at constant current. Also, transistors Q₅ and Q₆ are significant sources of distortion because the differential currents +I DO and -I DO vary as they flow through these transistors in generating the output voltage. This distortion can be made proportional to current by "bootstrapping" transistors Q₅ and Q₆ to force the collector-base junction of each transistor to be constant. Such bootstrapping is provided for transistor Q₆ by connecting its collector to one end of resistor R₅ (across which the voltage is constant) and its base to the other end. For transistor Q₅, bootstrapping is provided by transistors Q₇ and Q₈. Transistor Q₈ is tied to the supply voltage V CC and establishes the collector voltage of transistor Q₅. Transistor Q₇ is connected as a diode in the current path from the supply voltage V CC at the node between resistors R₄ and R₅. The base-emitter junction of transistor Q₇ steps the voltage at the node up one diode drop to the base of transistor Q₈. The base-emitter junction of transistor Q₈ then reduces the voltage one diode drop so that the voltage at the collector of transistor Q₅ equals the voltage applied at the node between resistors Q₄ and R₅. With the voltage across the collector-base junctions of transistors Q₅ and Q₆ held constant, thermal distortion within the transistors therefore varies linearly with the differential current flowing through the respective base-emitter junctions. This distortion is compensated for by thermal compensation means responsive to the input signal, such as the base-emitter junctions of the current source transistors Q₁₀ and Q₁₁. Transistor Q₁₀, for example, produces a distortion in current +I DO that causes the voltage across resistor R₂ to increase, raising the emitter voltage of transistor Q₅. Simultaneously, the increased heating of transistor Q₅ produces a distortion in current +I DO that decreases the emitter voltage a like amount. The thermal distortions are thus of opposite phase and can be made to offset each other. Resistor R₈ is a variable resistor or a resistor with a preselected resistance connected between the emitters of transistors Q₁₀ and Q₁₁ to vary the impedance and thus the amount of thermal distortion generated therein. As a variable resistor it may be adjusted, for example, to account for thermal distortion elsewhere in the circuit such as by the base-emitter junctions of transistors Q₉ and Q₁₂. FIG. 3 shows a second embodiment of the circuit 10 where the gain is doubled. Like reference symbols are retained in FIG. 3 for like components of FIG. 1. Resistors R₂ and R₃ have twice the value as they do in FIG. 1. Twice the PN junction compensation is required to double the voltage lost across the base-emitter junctions of transistors Q₁ and Q₂. Additional linearizing PN junctions of transistors Q₁₃ and Q₁₄ are included for this purpose. Resistors R₁₂ and R₁₃ are also included to compensate for α losses in transistors Q₁ through Q₄, making the gain of the converter less dependant on transistor α. Transistor Q₆ is split into two transistors Q 6A and Q 6B for isolating the bootstrapping of transistor Q₅ from the feedback to transistor Q₉. This change in the bootstrapping allows for greater freedom in establishing the DC operating conditions of the shunt feedback stage. Transistor Q 6B is flow biased by a separate means comprising supply voltage V CC and resistor R₁₁. Having illustrated and described the principles of the invention in preferred embodiments, it should be apparent to those skilled in the art that the invention can be modified in arrangement and detail without departing from such principles. For example, the present invention may be fabricated as an integrated circuit or as a discrete circuit.

Claims: A circuit for converting a differential input signal (Vin,-Vin) to a single-ended output signal (Vout), the circuit including an input stage (Q1,Q2) having PN junctions for receiving the differential input signal to produce a differential output current and having an output stage (Q12) to provide the single-ended output signal, being characterised by: means (Q3,Q4) coupled to the output terminals of the input stage for isolating the input stage to produce a buffered differential output current so that the voltage at the output terminals of the isolating means is isolated from the voltage at the input terminals of said means; PN junction means (Q5,Q6; Q5,Q6A,Q6B,Q13,Q14) matched to the PN junctions of the input stage and coupled to receive the buffered differential output current for applying to the output terminals of the isolating means a compensated differential voltage equal to the sum of a portion of the differential input signal lost across the PN junctions, and of a differential voltage which is proportional to the buffered differential output current, the input of the output stage being coupled to one of the two output terminals of the isolating means; and means (Vlin,R9,R10) coupled to the output terminals of the input stage for adjusting the compensated voltage by adjusting the differential current received at the input terminals of the isolating means so that the current through the PN junction means is adjusted. The circuit of claim 1 wherein the adjusting means comprises: a variable voltage source (Vlin); and a pair of resistors (R9,R10) coupled in series between the output terminals of the input stage with the variable voltage source coupled to the junction between the resistors, the variable voltage source being adjustable to adjust said differential current and thereby adjust the current drawn through the isolating means from the PN junction means. The circuit of claim 1 or 2 further comprising: means (Q9) coupled to the other one of said two output terminals of the isolating means (Q3,Q4) and coupled to the PN junction means for modifying thermal distortion generated by the PN junction means to vary linearly with the differential input signal; and means (Q10,Q11,R8) coupled to the input stage for generating responsive to the differential input signal thermal distortion of equal magnitude and opposite in phase to the thermal distortion of the PN junction means in order to cancel the thermal distortion of the PN junction means. The circuit of claim 3 wherein the generating means is adjustable to compensate for other sources of thermal distortion within the circuit which vary linearly with the differential input signal. The circuit of claim 3 or 4 wherein the generating means comprises: a pair of transistors (Q10,Q11) coupled to the input stage to provide current for the input stage, the transistors having PN junctions matching PN junctions of the PN junction means; and a resistor (R8) coupled between the emitters of the transistors. The circuit of claim 3, 4 or 5 wherein the PN junction means comprises a transistor (Q6) coupled to the input of the output stage to receive the one side of the differential output signal, and the modifying means is adapted to maintain a constant collector-base voltage across the transistor. The circuit of claim 1 or 2 comprising means (Q9) coupled to receive the other side of the differential output current for inverting and providing said side as a portion of the single-ended output signal contributed by the other side of the differential input signal, the difference portion between the two sides of the differential input signal being added to produce the single ended output signal and the common portion between the two sides of the differential input signal being rejected from the single-ended output signal. The circuit of claim 7 wherein the inverting means comprises a transistor coupled to the input stage via the isolating and PN junction means in a shunt feedback loop. The circuit of claim 7 or 8 further comprising means (R5) for maintaining a constant current into the inverting means to avoid the generation of error voltages in the single-ended output signal otherwise caused by a change in voltage across the inverting means in response to a current change through the inverting means.

Inventors: METZ ARTHUR J, METZ, ARTHUR J.
Applicants: TEKTRONIX INC, TEKTRONIX, INC.
Application year: 1988
Main classification: H03F3
IPC codes: H03F1, H03F3