Current-regulation of an error amplifier

Abstract: A deviation in the current gain characteristic of an error amplifier input transistor (Q148) from a nominal value which produces a corresponding deviation in its control electrode current, is compensated by the control electrode of another transistor (Q726) so as to substantially prevent such current deviation from affecting the output current of an input stage.

Description: The invention relates to a differential amplifier for a power supply, for example in a television apparatus. GB-A-2027306 discloses a differential amplifier having a pair of differential transistors the emitters of which are coupled to a current sink transistor of a first current mirror. An input at the base of one of the differential transistor is coupled to a second current mirror. A reference transistor which substantially corresponds to the said one of the differential transistor has a base coupled to the second current mirror and an emitter coupled to the first current mirror. The base bias current of the said one differential transistor and of the reference transistor is provided by the first mirror. The emitter current is provided by the second mirror. In this way the input signal source is not loaded by the base bias current. In accordance with the present invention, there is provided an amplifier comprising first and second transistors that are coupled to form a differential amplifier; an input stage coupled at a first junction terminal to a control electrode of said second transistor for establishing a voltage level at said control electrode of said second transistor in accordance with a voltage that is developed at an input terminal of said input stage such that an output current flowing in said input stage is coupled to said first junction terminal; a source of a first current; characterized by a third transistor; and a current mirror arrangement coupled to said third transistor and responsive to said first current and to a control electrode current of said third transistor for generating at an output terminal of said current mirror arrangement a second current that is coupled to said first junction terminal, said second current having a first portion derived from said first current and a second portion derived from said control electrode current of said third transistor, said current mirror arrangement supplies both a control electrode current of said second transistor and an output current of said input stage. In the drawing : FIGURE 1 illustrates a simplified schematic diagram of a power supply regulator circuit that includes an error amplifier and level shifter; FIGURE 2 illustrates a detailed schematic diagram of the error amplifier in accordance with the invention in combination with the level shifter of figure 1; and FIGURES 3, 4, 5 and 6 illustrate alternative level shifters. Figure 1 illustrates a simplified schematic diagram of a power supply of a television receiver, not shown in the figures, that includes a regulator 100 that is an integrated circuit that regulates a supply voltage B+. Voltage B+ may be used, for example, to energize a horizontal deflection circuit or output stage 99 of the television receiver. A voltage V+, representative of voltage B+, is obtained from output stage 99. Voltage V+ is coupled to a voltage divider 605 that includes series coupled resistors 601, 604 and 602. Resistor 604 includes a wiper k for developing at wiper K a voltage that is representative of, for example, voltage B+. The voltage at wiper K, that is adjustable by varying the position of wiper k, is coupled to an inverting input terminal 608 of an error amplifier 610 according to the present invention via a resistor 607. A small voltage that is proportional to the beam current in the CRT of the receiver is coupled from a tertiary winding of transformer T to a terminal 611 to form a voltage V NINI that is indicative of the beam current. Voltage V NINI that varies when a variation of the beam current occurs, is coupled via a level shifter 600, to a noninverting input terminal 609 of error amplifier 610 to produce an input voltage V NIN . Level shifter 600 establishes a fixed offset voltage between terminals 611 and 609 that is determined by a voltage V BG . Voltage V BG is generated in a bandgap type voltage source 699. Bandgap type voltage source 699 advantageously maintains voltage V BG constant such that voltage V BG is affected significantly less by component aging or tolerance than would have occurred had a zener diode been used. As explained later on the feed back arrangement of regulator 100 causes voltage B+ to be such that voltage V IN becomes equal to voltage V NIN . An integrating filter 612 is coupled between inverting input terminal 608 and an output terminal 618 of amplifier 610 to provide the loop filter of regulator 100. A filtered error voltage V o , developed at terminal 618 is coupled to a first input terminal of an adder 613. A horizontal rate sawtooth generator 98 develops a horizontal rate signal, having an upramping portion, which is added to error voltage V o in adder 613. The sum signal that is also upramping, is applied to an inverting input terminal 614 of a comparator 615 functioning as a pulse width modulator. When, during its upramping portion, the output of adder 613 becomes more positive than a constant DC voltage V REF , that is coupled to a noninverting input terminal of comparator 615, a negative going transition at an output terminal 615a of comparator 1615 is coupled via a buffer amplifier 616 to a control terminal 617a of a switch 617b of a switch mode power supply output stage 617 to turn on switch 617b of output stage 617. An input terminal 617c of output stage 617 is coupled to unregulated voltage V UR . Regulated voltage B+ is developed at an output terminal 617d of output stage 617. The duration, during each horizontal period, H, in which switch 617b conducts is determined by the level of error voltage V o of error amplifier 610. Thus, regulated voltage B+ is determined by voltage V NIN . As indicated before, voltage V NIN is produced by level shifter 600, embodying an aspect of the invention, that is described now in detail. FIGURE 2 illustrates a schematic diagram of level shifter 600 of FIGURE 1 and of error amplifier 610 according to the present invention. Similar numbers of FIGURES 1 and 2 represent similar items or functions. Level shifter 600 of FIGURE 2 is temperature compensated over a wide range of ambient operating temperatures, such as between 0°C and 70°C, to produce voltage V NIN that is substantially unaffected by a change in the temperature within such range. A temperature compensated current control arrangement 650 generates a control voltage V BR on a rail signal line 900. Rail signal line 900 is coupled to the base electrode of each of transistors Q142, Q725, Q727, Q736 and Q737. The emitter electrodes of the abovementioned transistors are coupled through corresponding resistors to a fixed DC voltage V cc . Transistors Q725 and Q727 have their emitters both connected to source resistor R727. Current control arrangement 650 controls voltage V BR in such a way that the collector current in each of the abovementioned transistor e.g. Q727, Q725 stays substantially constant when the temperature changes. An example of an arrangement that is similar to current control arrangement 650 is described in detail in US Patent No. 3,886,435, in the name of S.A. Steckler, entitled VBE VOLTAGE SOURCE TEMPERATURE COMPENSATION NETWORK. Level shifter 600 includes transistors Q736 and Q737. The emitter currents in transistors Q736 and Q737 are controlled by resistors R728 and R69, respectively, having the same value, so as to cause the respective collector currents of transistors Q736 and Q737, that are temperature compensated, to be equal. The collector of transistor Q737 is coupled to a current mirror arrangement that includes transistors Q733, Q734 and Q735. The collector of transistor Q737 is coupled to the collector of transistor Q734. The emitter of transistor Q735 is coupled to each of the bases of transistors Q733 and Q734. Transistor Q735 provides the base current drive to each of transistors Q733 and Q734. The emitter of transistor Q734 is coupled to ground via a resistor R732. The P-N junction of transistor Q734 between the base and emitter electrodes of transistor Q734, provides temperature compensation that compensates for a temperature related variation of the base-emitter voltage of transistor Q733. The emitter of transistor Q733 is coupled through resistor R731 to terminal 611, where voltage V NINI of FIGURE 1 is developed. The value of resistor R731 is equal to that of resistor R732. Voltage V NINI is prevented from exceeding predetermined limits in either polarity by a diode network 675. The collector of transistor Q733 is coupled to the collector of transistor Q736 at a junction terminal 733A. Assume that voltage V NINI is zero. In this case the current mirror arrangement of transistors Q733, Q734 and Q735, produces a collector current i Q733 in transistor Q734 because the base current of transistor Q735 is negligible. As explained before, when voltage V NINI is zero, collector current i Q736 in transistor Q736 is equal to collector current i Q737 in transistor Q737 over a wide temperature range. Also, when voltage V NINI is zero, each of collector current i Q734 over such wide temperature range. It follows that current i Q733 is also equal to current i Q736 . Bandgap type voltage source 699 supplies temperature compensated reference voltage V BG that is coupled via a resistor R729 to terminal 733A. Because, as described before, when voltage V NINI is zero, current i Q733 is supplied entirely by current i Q736 , and because the impedance at terminal 733A, that is contributed by the collectors of transistors Q733 and Q736 is high, a current i R729 in resistor R729 is zero; therefore, voltage V NIN at terminal 733A is equal to voltage V BG . Thus in accordance with an aspect of the invention, when voltage V NINI is zero, voltage V NIN is level shifted by an amount that is equal to voltage V BG . When voltage V NINI at terminal 611 is different from zero, currents i Q736 and i Q737 will not be equal. The difference current between currents i Q733 and i Q736 will cause a voltage to develop across resistor R729 that, in turn, will cause a corresponding change in voltage V NIN at terminal 733A. Because transistor Q733 is coupled, relative to voltage V NINI , as a common base amplifier, and because resistors R731 and R729 are, illustratively, equal, the gain, or the ratio between voltage V NIN and voltage V NINI , is one, resulting in an amplifier having a unity gain. Voltage V NIN , that is level shifted relative to voltage V NINI by an amount that is equal to voltage V BG , follows variations of voltage V NINI that occur in a range between positive and negative values. Voltage V BG is temperature compensated and has a tolerance range that is narrow relative to, for example, a zener diode. Furthermore, component aging affects voltage V BG substantially less than it affects, for example, the breakdown voltage of a zener diode. Moreover, the level shifting caused by level shifter 600 is, advantageously, less susceptible to temperature, aging and noise when compared with that produced by a corresponding level shifter in the prior art that utilizes a zener diode interposed between a beam current input terminal and a noninverting input terminal of a differential amplifier to perform such level shifting. Should a temperature change cause a corresponding change in current i Q736 , for example, that, as indicated before, would be relatively small, transistors Q737, Q733, Q734 and Q735 will cause a proportional change in current i Q733 to occur that will prevent even such small change in temperature from affecting the difference current between currents i Q736 and i Q733 . Therefore, when voltage V NINI is zero, voltage V NIN is, advantageously, not affected by collector currents i Q736 and i Q737 , but is entirely determined by voltage V BG that is temperature compensated. It should be understood that temperature compensation may be adequate even when voltage V NINI is significantly different from zero. If temperature compensation, in this case, is inadequate, a further improvement in temperature compensation may be obtained by coupling the terminal of, for example, resistor R732, that in FIGURE 2 is grounded, to a voltage that is different from zero and that is related to, for example, voltage V NINI . Advantageously, voltage V BG , as explained before, is maintained at tight tolerances, is temperature compensated and is substantially unaffected by components aging. Therefore, advantageously, no factory temperature burn-in process is required prior to the instalment of regulator 100 of FIGURE 1 in the television receiver. Furthermore, voltage divider 605 that includes resistors 601, 604 and 602 is required to compensate, advantageously, only for a narrower tolerance range than in prior art circuits in which a zener diode is used for performing the level shifting function of level shifter 600 of FIGURE 2. The error amplifier 610 according to the present invention will now be described in more detail. As shown in Figure 2, the error amplifier 610 includes a differential amplifier comprising transistors Q148 and Q149, the emitters of which are coupled to a common current source transistor Q142. Voltage V in at terminal 608 is coupled to the base of transistor Q148 via an emitter-follower transistor Q721. Voltage V NIN at terminal 609 is coupled to the base of transistor Q149 via an emitter-follower transistor Q723. The clamping operation of a pair of transistors Q145 and Q146 prevents voltage V IN from being above voltage V BG or from being below voltage V BG by more than a predetermined magnitude. A first current mirror arrangement 610b that is coupled to the collectors of transistors Q148 and Q149 causes a current i 610 , coupled to integrating filter 612 of FIGURE 1, to be equal to the difference between the collector currents in transistors Q148 and Q153. Consequently, current i 610 that is coupled to filter 612 of FIGURE 1, is proportional to the difference between voltages V IN and V NIN . The proportionality factor is determined by the gain of error amplifier 610. A second current mirror arrangement is coupled to the bases of the differential transistors Q148, Q149. The second arrangement comprises like transistors Q722, Q724 having collectors coupled to the bases, emitters coupled by like resistors R724, R725 to ground and bases commonly coupled to the collector and base of a mirror transistor Q730. The emitter of transistor Q730 is coupled to ground via resistor R726 equal to resistors R724 and R725. The collector base connection of mirror transistor Q730 is connected to the collector of transistor Q727 and via a transistor Q726 to the collector of transistor Q725. The base of transistor Q726 is connected to mirror transistor Q730 and the collector emitter path of Q726 is connected to ground and to the collector transistor Q725. The total current flowing through Q730 and thus V BE of Q730 is determined by the sum of the collector current of Q727 and the base current of Q726. V BE of transistors Q722 and Q724 is determined by V BE of mirror transistor Q730, and thus the collector emitter currents of transistor Q722 and Q724. The collector emitter currents of transistor Q722 (or Q724) equals to the sum of the base current of transistor Q148 (or Q149) and the emitter current of transistor Q721 (or 723). As discussed above controller 650 maintains the collector current of Q727 constant with varying temperature. FIGURES 3, 4, 5 and 6 illustrate alternative level shifters 600a, 600b, 600c and 600d, respectively. In FIGURES 2-6, numbers and symbols of similar items or functions are similar except that they include the letters a, b, c and d, in FIGURES 3, 4, 5 and 6, respectively. In FIGURE 3, resistors R731 and R732, that are used in the circuit of FIGURE 2, were eliminated. Without resistors R731 and R732, the gain of amplifier 600a of FIGURE 3 is, advantageously, higher than unity. In FIGURE 3, a diode D cma and a temperature compensated current source I 1a cause the collector currents in transistors Q737a and Q736a to be, for example, equal. Similarly, transistor Q734a causes the collector current in transistor Q733a to be, for example, equal to that in each of transistors Q737a and Q736a when voltage V NINI is zero. In FIGURE 4, voltage VNINIb is applied differentially between the emitters of transistors Q733b and Q734b. Such arrangement provides, advantageously, an improved common mode rejection. In FIGURE 5 the type N-P-N or P-N-P, of the corresponding transistors is opposite than that in FIGURE 2 so that voltage V NINIc may, if desired, be referenced to voltage V cc , instead of to ground. In FIGURE 6, the input impedance to voltage V NINId is, advantageously, higher than to voltage V NINI of FIGURE 2 because of the usage of a transistor Q750 that is coupled as an emitter follower.

Claims: An amplifier comprising : first (Q149) and second (Q148) transistors that are coupled to form a differential amplifier (610); an input stage (Q721) coupled at a first junction terminal (base of Q148) to a control electrode (base) of said second transistor (Q148) for establishing a voltage level at said control electrode of said second transistor (Q148) in accordance with a voltage (V in ) that is developed at an input terminal (608) of said input stage such that an output current flowing in said input stage is coupled to said first junction terminal; a source (Q727) of a first current; characterized by a third transistor (Q726); and a current mirror arrangement (Q722,Q730) coupled to said third transistor (Q726) and responsive to said first current (from Q727) and to a control electrode current of said third transistor (Q726) for generating at an output terminal of said current mirror arrangement (Q722,Q730) a second current (at collector of Q722) that is coupled to said first junction terminal (base of Q148), said second current having a first portion derived from said first current (from Q727) and a second portion derived from said control electrode current of said third transistor (Q726), said current mirror arrangement supplies both a control electrode current (i bQ148 ) of said second transistor and an output current (I eQ72l ) of said input stage. An amplifier according to Claim 1, characterized in that said input stage comprises a fourth transistor (Q721) coupled as am emitter follower, and said output current of said input stage flows in an emitter electrode of said fourth transistor (Q721). An amplifier according to Claim 1, characterized in that said first current source (Q727) generates said first current having a constant value. An amplifier according to Claim 1, characterized in that said first (Q149), second (Q148) and third (Q726) transistors are bipolar transistors of the same type. An amplifier according to Claim 1, characterized in that said current mirror arrangement (Q722,Q730) includes a fourth transistor (Q722) that generates said second current that flows in the same direction as said first current (in Q727) and that is equal to the sum of said first current and said control electrode current of said third transistor (Q726). An amplifier according to Claim 1, characterized in that said third transistor (Q726) is coupled outside a signal path formed between an input terminal (608) and an output terminal (610a) of said amplifier. An amplifier according to Claim 1, characterized by means (650) for generating a temperature compensated control voltage (V BR ) that is coupled to a control electrode of a bipolar, further transistor (Q727) for generating said first current at a collector electrode of said further transistor (Q727) that is unaffected by a variation of a current gain characteristic of said further transistor (Q727). An amplifier according to Claim 7, characterized in that said current mirror arrangement further comprises an additional transistor (Q730) coupled in a diode configuration for conducting therethrough a current that is equal to a sum of said first current (from Q727) and said control electrode current of said third transistor (Q726). An amplifier according to Claim 8, characterized in that said current mirror arrangement includes another transistor (Q722) having a base electrode that is coupled to said additional transistor (Q730) and a collector electrode that is coupled to said control electrode of said second transistor (Q148). An amplifier according to Claim 9 further comprising a second additional transistor (Q724) having a base electrode that is coupled to said additional transistor (Q730) and a collector electrode that is coupled to a base electrode of said first transistor (Q149). An amplifier according to Claim 1, characterized in that said amplifier forms an error amplifier in a regulator of a television apparatus power supply.

Inventors: CRAFT JACK, LOW MICHAEL LOUIE, YORKANIS BERNARD JOSEPH, CRAFT, JACK, LOW, MICHAEL LOUIE, YORKANIS, BERNARD JOSEPH
Applicants: RCA THOMSON LICENSING CORP, RCA THOMSON LICENSING CORPORATION
Application year: 1987
Main classification: G05F3
IPC codes: G05F3, H04N5