A CIRCUIT FOR BIASING A MAGNETORESISTIVE SENSOR AND AMPLIFYING SIGNALS PRODUCED THEREBY

Abstract: A constant current source (J1) supplies a bias current to a magnetoresistive (MR) sensor (R h ) and amplifier means (11, 12, 13) is responsive to the current flowing through the MR sensor to produce an amplified output signal representing the change in resistance of the MR sensor arising from changes in the magnetic field to which the MR sensor is exposed. The circuit includes first means (14) coupled to the amplifier means to set the frequency response thereof and second means (15) for substantially eliminating DC offset voltage in the output signal. The resistance of the MR sensor can be disposed as degenerative feedback in the emitter circuit of a differential pair comprising the input stage of the amplifier. The constant current source (J1) that supplies bias current to the MR sensor also supplies current to the differential pair. DC feedback to the input stage balances current flow in both paths of the differential input stage to correct for DC offset arising in the output from variations in transistor characteristics and the steady-state value of the MR sensor.

Description: This invention relates to a circuit for biasing a magnetoresistive sensor and amplifying signals produced thereby. Most existing magnetoresistive (MR) sensors, including particularly sensors for bubble memories and magnetic recording, are biased at a constant bias current by preamplifiers which detect the signal voltage developed at the sensor terminals. Since stripe height is inversely proportional to sensor resistance, and varies from sensor to sensor owing to production variations and varies over the life of the sensor in contact recording applications owing to wear, it is desirable to detect signals produced by the sensor which are independent of stripe height. Thus, it is desirable to detect and amplify a voltage representing ▵ R h /R h , where ▵ R h is the change in MR sensor resistance, R h, arising from the magnetic input signal from the media being read. Prior art document EP -A- 215270 relevant under Art.54(3) EPC, discloses a circuit comprising a bias current source coupled to a magnetoresistive element to produce a current therethrough having a constant time average value and means for detecting and amplifying the instantaneous value I S of current deviations from the constant time average value substantially in accordance with the relationship: where I b is the current provided by the bias current source, R h is the absolute resistance of the magnetoresistive element, and ▵ R h is the instantaneous value of the change in the resistance of the magnetoresistive element in the presence of the magnetic field. DE-A-33 08 352 from which the preamble of present Claim 1 derives describes magnetic detection apparatus comprising a first magnetoresistive sensor exposed to an external magnetic field and a second magnetoresistive sensor exposed to both the external magnetic field, and a desired signal magnetic field. A differential amplifier means is provided with a negative feedback loop for feeding back an output signal from the first sensor so as to cancel the external magnetic field component applied to the second sensor. The differential amplifier means thereby enables the signal magnetic field to be detected separately from any external magnetic fields. The application of negative feedback to differential amplifier circuits for preventing transient instability is well known in the art. It is also well known that many such negative feedback techniques reduce the frequency response bandwidth of such amplifiers. In accordance with the present invention, there is now provided a circuit comprising a first constant current source to supply a bias current to a magnetoresistive sensor, and amplifier means having an input stage responsive to the current flowing through the sensor to produce by output means an amplified output signal representing the change in resistance of the sensor arising from the changes in the magnetic field to which the sensor is exposed, and feedback means, characterised in that the feedback means are coupled to the input stage and the output means for adjusting the frequency response of the circuit, wherein the input stage comprises at least two transistors coupled in a first differential pair configuration and having the sensor coupled between the emitter of one of the pair of transistors and the first constant current source, the emitter of the other transistor also being coupled to the first constant current source, and in that the output means comprises at least two transistors coupled in a second differential pair configuration, the respective collectors of these transistors being coupled to the respective bases of the transistors of the first differential pair for correcting voltage level variations at the bases of the transistors of said first differential pair, and first and second follower transistors coupled to the respective collectors of the transistors of said first differential pair, for coupling voltage level variations produced in the input stage to the second differential pair, thus substantially eliminating any DC offset voltage in the amplified output signal. In a preferred embodiment of the present invention, an enhanced transimpedance amplifier is used for biasing and amplifying the signals produced by an MR read sensor. Electrically, the resistance, R h , of the MR sensor is disposed as degenerative feedback in the emitter circuit of a differential pair comprising the input stage of the amplifier. In this circuit configuration, a signal representing ▵ R h /R h is sensed and amplified as a current through the MR sensor. Bias current for the MR sensor is supplied by the same constant current source that supplies current to both transistors comprising the input stage of the amplifier. To correct for direct current (DC) offset arising from variations in input stage transistor characteristics and the steady-state value of R h , DC feedback to the input stage, via a level shift and amplifying stage, balances current flow in both paths of the differential input stage. It should be noted that the steady-state value of R h refers to the resistance value of R h while biased but not sensing magnetic field variations. In addition, a feed forward current source, which injects an offset current into the last-mentioned DC feedback loop, may be used to correct for the DC error at the output of the amplifier. The amount of current can be selected to eliminate the error for nominal values of steady-state sensor resistance R h , and to minimise the error in the general case. It should be noted that nominal, in regard to the value of R h , refers to that steady-state value of R h most often occurring in production quantities. The input stage of the amplifier may be replicated as many times as there are MR sensors in a given system. The remaining components of the amplifier comprise the output and feedback stages which are common to all, and switched among, the input stages in a multiple-head system. The sensitivity of the amplifier to ▵ R h /R h signals may be increased and the sensitivity to variations in R h may be decreased by additional circuitry. Further additional circuitry may be added to shorten settling time when MR sensors having different steady state resistance values, R h , are activated and deactivated in a multiple sensor system. How the invention can be carried out will now be described by way of example, with reference to the accompanying drawings, in which:- Fig. 1 is a block diagram of a circuit embodying the invention; Fig. 2A is a circuit diagram corresponding to the block diagram of Fig. 1; Fig. 2B is a block diagram indicating the coupling of input stages to the common stage of the amplifier; Fig. 3 is a circuit diagram for enhancing ▵ R h /R h sensitivity of the amplifier in Fig. 2A; Fig. 4 is a circuit diagram for improving transient response of the amplifier in Fig. 2A when switched between MR sensors having different resistance; Fig. 5 is a block diagram of elements for reducing the settling time of the amplifier in Fig. 2A when MR sensors having different resistance are activated and deactivated in a multiple-sensor system; Fig. 6 is a circuit diagram corresponding to Fig. 5; and Fig. 7 is a diagram of a circuit emboding the enhancements of Figs. 4 and 6. Referring first to Fig. 1, input I from an MR sensor is applied to summer 11 of amplifier 10. Since ▵ R h is proportional to the resistance value Of R h , the output signal can be made independent of variations in the value of R h arising from variations in stripe height and length by dividing the total gain of the system by R h . Thus, the forward transfer function of gain stage 12 is G divided by R h . The reverse transfer function, H, of feedback network 14 can be modified by setting dominant low frequency poles, Pd, in a low pass configuration to produce a high pass characteristic of the system at the output, O. The high pass characteristic is obtained by subtracting error signal, E, produced by network 14, from the input, I, at summer 11. Finally, feed forward correction, F, from source 15 for correcting DC offset at output 0 is introduced at summer 13. The input I contains an undesirable DC component arising from the voltage drop across the sensor caused by the bias current. This DC component may create an error at output O. Therefore, subtracting the nominal value of the DC component in I from the input to the forward gain Path (G/R h ) eliminates the error in output O for the nominal values of R h , and significantly reduces the error in output O for the general case of R h values at or near the tolerance limit of acceptable values thereof. The concept of Fig. 1 may be implemented as shown in Fig. 2A. Bias current, I b , for the MR sensor, R h , is supplied by current source J1. Initially, the current through input devices T1 and T2 may be different and will cause an output offset across R1 and R2. Voltage followers T5 and T6, each coupled to constant current sources J3 and J4, respectively, shift the DC potential of this offset and apply it to feedback differential pair, T7 and T8. Thus, the voltage at the bases of input devices T1 and T2 is corrected so that one-half of the current from current source J1 flows in each input device T1 and T2. The current supplied by constant current source J1 is approximately 2I b . Differential pair, T7 and T8, are supplied with current by constant current source J2. Since input devices T1 and T2 have inherently low input impedance at their emitters, the circuit is a transimpedance amplifier. In addition, R h acts as a degenerative feedback resistor. Thus, ▵ R h/ R h is sensed as a current through R h . Referring now also to Fig. 2B, nodes n1, n2 and n3 and NF1 and NF2 define common nodes joining separate circuits. The first circuit, comprising input devices T1 and T2, R h and current source J1, is the basic input stage for each MR sensor. The remaining components shown in Fig. 2A comprise the output and feedback stages, hereafter collectively referred to as g 0 , which are common to two or more basic input stages. Thus, each MR sensor, R h1 , R h2 ... R hn , has a separate input stage, g₁, g₂, ... g n ' which can be activated or deactivated by activating or deactivating their respective current sources, J1. With continuing reference to Fig. 2A, capacitor C, coupled between the bases of T1 and T2, is used to produce a dominant low frequency pole as indicated in Fig. 1 The location of the pole is determined by the value of C, R5, R6 and the loop gain, i.e. GH/R h . Current source J0 injects a feed-forward offset current into the feedback loop to balance the collector voltages of output cascade pair T3 and T4 for nominal MR resistance, R h , and minimises such output offset in the general case for a range of steady-state values of R h . The values of resistors R1, R2, R3, R4, R5, R6, R7 and R8 are selected to provide suitable voltage biasing for their respective active devices. Furthermore, the circuit of Fig. 2A is particularly well-suited to implementation in any number of different integrated circuit technologies, since only one capacitor, which could be an off-chip component, is required. The output voltage, V out , of the amplifier is related to the input impedance, R in , as seen by R h as given below, Therefore, Thus, the lower the value of R in , the less sensitive V out will be to a wider range of variation sin the steady-state value of R h . For assuring very low input impedance, R in , presented to the MR sensor, R h , the amplifier may include the compound input stage of Fig. 3. The circuit of Fig. 3 includes additional current sources J5 and J6, which are common to all input stages. Transistors T₁₁, T₁₂, T₉, T₁₀, and diodes D ₁ and D ₂ are all repeated for each input stage. Then, additional nodes separating the common from the repeated parts of the circuit are identified at N4, N5, N6 and N7. Wide-band performance for the circuit of Fig. 2A is desirable and obtainable by 1) setting the dominant low frequency pole for good low frequency response, and 2) by adding a cascaded common-base stage comprising T ₃ , T ₄ and V₁ for good high frequency response. However, too much recovery time is required for stabilising such a circuit when switching over from one sensor to another, i.e. deactivating one sensor and associated input stage and activating another. Obviously, if the forward gain G were increased, the dominant low frequency pole would move up in frequency and recovery time would decrease. Since that solution is undesirable for overall performance, a circuit is included for moving the dominant low frequency pole only during switchover. The circuit for Fig. 4 comprises a non-linear feedback loop which improves the transient response of the circuit of Fig. 2A when successively switching over input stages having different values of R h. Diodes D9 and D10, resistor R₇ and current source J9 comprise additions to the common output/feedback stage of Fig. 2A. These last-mentioned components produce higher gain for reverse transfer function network 14, shown in Fig. 1, for large error signals. In particular, as the impedance of diodes D9 and D10 decreases with increasing current therethrough, the degenerative resistance in the emitter circuit of the differential pair T7 and T8 is further reduced for large input signals. Thus, the dominant low frequency pole, Pd, tends to move up in frequency for a faster transient response in accordance with the relation O/I = [(1/Ho)(s+Pd)]/[R h /GHo)(s+Pd)+1] (1) where Ho=DC gain of the reverse path of the feedback loop, s=Laplace operator, Pd=dominant pole, R h =sensor resistance and G=the uncorrected gain of the forward path of the feedback loop. In accordance with Equation (1), as the error signal approaches the steady state value, the low frequency pole tends to move back to the initial frequency for good low frequency response at output 0. Fig. 5 illustrates the principle of a non-linear feedback loop for reducing the time required for the amplifier to settle into steady-state operation after one input stage is deactivated and another is activated. The non-linear feedback loop is active only during part of the transition time when switching between input stages. While active, this feedback loop introduces additional current to capacitor C to reduce the relatively long charge and discharge times thereof, which current is otherwise limited by its value and the charging current supplied by current source J2 as shown in Fig. 2A. With continuing reference to Figs. 2B and 5, when R h1 is biased for steady-state operation, the voltage across capacitor C is fixed and maintained by feedback. If R h1 is then deactivated and R h2 is activated, where R h2 is greater than R h1 , then a DC offset voltage appears at the output, V ₀ . Employing preselectable hysteresis in comparator CM2, if V₀ then exceeds V ref by V on , then comparator CM2 activates current source J11. Capacitor C is quickly charged to its new bias voltage because current source J11 temporarily provides substantially greater current than the charging current of G₁ (i.e., current source J2 as shown in Fig. 2A). When the new steady-state operating voltage is achieved, the DC offset at V₀ is decreased. When the DC offset V ₀ is decreased below V off , which is less than V on , then comparator CM2 deactivates current source J11 and the linear feedback loop comprising G₁, G₂ and the input stage further reduces any residual offset at V₀. Likewise, if sensor R h2 is switched off and new sensor R h3 is switched on, where R h3 is less than R h2 , comparator CM1 will operate to discharge capacitor C with current source J10. The voltage level V on is selected so that noise or other device parameter variations do not cause activation of comparators CM1 or CM2 while, at the same time, allowing operation of the comparators for small differences between activated and deactivated MR sensors. The voltage level V off is selected so that the comparators will not switch off too soon to provide minimum transient response of the amplifier while providing an adequate margin of loop stability. Referring now to Fig. 6, the basic input and common output feedback stages of Fig. 3 are coupled to an implementation of the non-linear feedback loop of Fig. 5. Separate reference voltages are derived from resistor dividers R22/R23 and R24/R25 to minimise interaction between the two comparators. Comparator CM2 comprises transistors T28, T30, T14, T16, T18 and T20. Diode D12 is used as a voltage level shifter. During steady-state operation T28 is kept on (i.e. actively conducting current) by T14. The voltage developed across R20 is V on . As the voltage at the base of T28 decreases and T30 turns on, T30 is kept on by T20. The voltage developed across R22 and R23 from T20 is equivalent to V off . Diode D4 is used as a voltage level shifter for T8. When the non-linear feedback loop is activated, transistor T22 supplies current to diodes D6 and D8. The base of T24 and thus, one side of capacitor C is thereby clamped to a fixed voltage. At the same time, T26 will charge capacitor C toward the new steady-state voltage level in minimum time. Since diodes D6 and D8 represent a high impedance when current through them is small and a low impedance when through them is large, the feedback loop is inherently non-linear for decreasing MR sensor switchover recovery time. A circuit for moving the dominant low frequency pole up in frequency while concurrently increasing available charge and discharging current to capacitor C is shown in Fig. 7. The circuit comprises a transconductance amplifier in a feedback loop in lieu of comparators, voltage reference levels and selectable voltage differences required by the circuits of Figs. 4 and 6. In the circuit of Fig. 7, the current supplied from current source J12 is set to a low value for steady-state operation. the current, in turn, sets the value of the current flowing in transistors T7, T8, T32 and T33 to a value I c1 . Current I c1 also flows through transistors T15 and T16 and transistors T13 and T14 and therefore becomes the current available to charge capacitor C. If I c1 were increased to a value I ch at the moment of switchover, then both the available charge current to capacitor C and the loop gain would increase, thereby reducing capacitor charge time and moving the dominant low frequency pole up in frequency, respectively. After sufficient time for recovery has passed, the current supplied by current source J12 may be decreased to the initial value, I c1 , for steady-state operation. The rate of increase of current I c1 to I ch can be controlled for smooth transition between switchover operation and steady-state operation of the circuit.

Claims: A circuit comprising a first constant current source (J1) to supply a bias current to a magnetoresistive sensor (R h ), and amplifier means (11,12,13,14) having an input stage (11,13) responsive to the current flowing through the sensor (R h ) to produce by output means an amplified output signal representing the change in resistance of the sensor (R h ) arising from the changes in the magnetic field to which the sensor (R h ) is exposed, and feedback means (14), characterised in that the feedback means (14) are coupled to the input stage (11,13) and the output means (12) for adjusting the frequency response of the circuit, wherein the input stage (11,13) comprises at least two transistors (T1,T2) coupled in a first differential pair configuration and having the sensor (R h ) coupled between the emitter of one (T2) of the pair of transistors (T1,T2) and the first constant current source (J1), the emitter of the other transistor (T1) also being coupled to the first constant current source (J1), and in that the output means (12) comprises at least two transistors (T7,T8) coupled in a second differential pair configuration, the respective collectors of these transistors (T7,T8) being coupled to the respective bases of the transistors (T1, T2) of the first differential pair for correcting voltage level variations at the bases of the transistors (T1,T2) of said first differential pair, and first and second follower transistors (T5,T6) coupled to the respective collectors of the transistors (T1,T2) of said first differential pair, for coupling voltage level variations produced in the input stage (11,13) to the second differential pair, thus substantially eliminating any DC offset voltage in the amplified output signal. A circuit as claimed in Claim 1, wherein the input stage (11,13) comprises means (11) for subtracting an error signal produced by the feedback means (14) from a signal representing the current flowing through the sensor (R h ). A circuit as claimed in Claim 1 or Claim 2, wherein the second differential pair feeds forward a signal to the input stage (11,13). A circuit as claimed in any preceding claim, wherein the input stage comprises means for dividing the signal representing the current flowing through the sensor (R h ) by a signal representing the steady state resistance of the sensor (R h ). A circuit as claimed in Claim 1 wherein the current provided by the first constant current source (J1) has a constant time average value, and the transistors (T1,T2) of the first differential pair amplify current deviations from the constant time average value of the current flowing through the sensor (R h ). A circuit as claimed in Claim 5, wherein said second differential pair (T7,T8) is coupled to a second constant current source (J2), and said follower transistors (T5,T6) are coupled to third and fourth constant current sources (J3,J4) respectively. A circuit as claimed in Claim 6, including a plurality of input stages (g1,g2) coupled to selection means for activating an individual one of the plurality of input stages and deactivating all of the remaining input stages. A circuit as claimed in Claim 7, including transient suppression means for suppressing transients produced by deactivating one and activating another of the plurality of input stages. A circuit as claimed in Claim 7 or Claim 8, wherein the transient suppression means comprises: a fifth constant current source (J9) coupled to the emitter of one of said differential pair (T7,T8); and a resistor (R7) and first and second diodes (D9,D10), each coupled in parallel to each other and between the second (J2) and a fifth (J9) constant current source, the cathode of the first diode (D10) and the anode of the second diode (D9) being coupled to the fifth constant current source (J9). A circuit as claimed in any of Claims 7 to 9, further including enhancing means (J5,J6) for decreasing input impedance presented to the sensor (R h ) and for increasing input impedance of the input stage. A circuit as claimed in Claim 7 or 8, further including settling means (D9,D10,R7) for reducing the time required for the circuit to settle into steady state operating after one of the input stages is deactivated and another of the input stages is activated.

Inventors: JOVE, STEPHEN ALAN, KLAASSEN, KLAAS BEREND
Applicants: INTERNATIONAL BUSINESS MACHINES CORPORATION
Application year: 1987
Main classification: G01R33
IPC codes: G11B19, G01R33, G11B5