DELAY CIRCUIT COMPRISING ALL-PASS NETWORKS

Abstract: A third-order all-pass network for a delay circuit is formed by four coupled transconductors (G₁₀ to G₁₃) which are each represented by two transistors whose bases constitute the inputs and whose collectors constitute the outputs of the transconductor. A first input of these transconductors (G₁₀ to G₁₃) is connected to earth (3). Between the second inputs (25,26) of the first transconductor (G₁₀) and the second transconductor (G₁₁) a first capacitor (C₁) is arranged, between the second inputs (26,27) of the second transconductor (G₁₁) and the third transconductor (G₁₂) a second capacitor (C2) is arranged, and between the second inputs (27,28) of the third transconductor (G₁₂) and the fourth transconductor (G₁₃) a third capacitor (C3) is arranged. Further, a fourth capacitor (C4) is arranged between the second inputs (25,27) of the first transconductor (G₁₀) and the third transconductor (G₁₂), a fifth capacitor (C5) is arranged between the second inputs (26,28) of the second transconductor (G₁₁) and the fourth transconductor (G₁₃), and a sixth capacitor (C6) is arranged between the second inputs (25,28) of the first transconductor (G₁₀) and the fourth transconductor (G₁₃). The third (C3), fourth (C4) and sixth (C6) capacitors reduce resonance tendencies of the circuit and consequent resonant rise symptoms as a result of parasitic effects.

Description: The invention relates to a delay circuit comprising at least one all-pass network which has an input terminal for receiving an input signal, an output terminal for supplying an output signal, and a reference terminal for carrying a reference potential and which comprises an at least second-order filter section comprising three transconductors, which each have a first input, a second input, a first output and a second output, the first input of each of the three transconductors being coupled to the reference terminal, the second input of the first transconductor being coupled to the input terminal, the second input of the second transconductor being coupled to the second output of the first transconductor and to the first output of the third transconductor, the second input of the third transconductor being coupled to the second output of the second transconductor and to the output terminal, which filter section further comprises a first capacitor arranged between the second inputs of the first transconductor and the second transconductor, and a second capacitor arranged between the inputs of the first transconductor and the third transconductor. Such a delay circuit may be employed for envelope-delay correction of filters, as a delay line whose delay may be variable or not, and in adaptive filters. Such a circuit arrangement is known from the article "Integration of Analog Filters in a Bipolar Process" IEEE Jnl. of Solid State Circuits, Vol. SC-17, No. 4, August 1982, pp. 713-722. A transconductor is a voltage-controlled current source in which the proportionality factor between the output current and the input voltage is given by the transconductance. By means of a transconductor it is possible to simulate a resistor, whilst two coupled transconductors may be arranged to form a gyrator by means of which a capacitor and an inductor can be simulated. This means that using transconductors and capacitors it is possible to realise all filter circuits which can also be manufactured by means of conventional coils, capacitors and resistors. From the above article it is also known that all-pass networks can be formed by means of transconductors and capacitors. Fig. 15 in the above article shows a first-order network and a second-order network comprising these elements. The transconductors have one input connected to earth and all the capacitors are floating. For the delay of signals of comparatively large bandwidth at least second-order all-pass networks are required in order to maintain deviations from the desired delay time comparatively small over the entire bandwidth. A drawback of such second-order all-pass networks is that parasitic effects, for example, as the result of stray capacitances, may show resonance effects. The gain of the all-pass network is then not equal to unity over the entire bandwidth of the signal and comparatively large frequency-dependent deviations from the desired delay occur. Therefore, it is the object of the present invention to provide a delay circuit provided with at least one all-pass network in which the parasitic effects show less resonance effects are reduced. According to the invention a delay circuit of the type defined in the opening paragraph is characterized in that the filter section comprises a third capacitor which is arranged between the second inputs of the second transconductor and the third transconductor. This third capacitor reduces the resonant effects, so that it is better possible to maintain unity gain over the entire bandwidth of the signal to be delayed and deviations from the desired delay are minimised. An embodiment of the invention may be characterized in that the all-pass network further comprises a first-order filter section which together with the second-order filter section constitutes a third-order filter section, which first-order filter section comprises a fourth transconductor having a first input and a second input and a first output and a second output, and a fourth capacitor. By constructing the all-pass networks as third-order networks deviations from the desired delay can be reduced relative to those occurring in second-order networks. This embodiment enables the likelihood of the occurrence of the resonance effects to be reduced even further if it is characterized further in that the fourth transconductor has its first input coupled to the reference terminal, its second input to the second output of the third transconductor, its first output to the second input of the third transconductor, and its second output to the output terminal of the network, in that the fourth capacitor is arranged between the second inputs of the third transconductor and the fourth transconductor, and in that the third-order filter section further comprises a fifth capacitor which is arranged between the second inputs of the second transconductor and the fourth transconductor and, at option, a sixth capacitor which is arranged between the second inputs of the first transconductor and the fourth transconductor. Deviations from the desired delay can also be reduced if the all-pass network does not comprise one but a cascade of at least two second-order filter sections, the third transconductor of the first section and the first transconductor of the second section being formed by a single transconductor which is common to both sections. If the delay circuit in accordance with the invention comprises a plurality of cascaded all-pass networks it may be characterized, in accordance with a further embodiment, in that every transconductor of an all-pass network which is connected to an output terminal and the transconductor of a subsequent all-pass network which is connected to the input terminal are formed by a single transconductor which is common to both sections. A further embodiment of the invention may be characterized in that the first output of the transconductor of the first all-pass network which is connected to the input terminal is connected to the reference terminal. Yet another embodiment may be characterized in that the output terminal of the last all-pass network of the delay circuit is connected to the second output of the transconductor connected to the output terminal. In the delay circuit in accordance with a further embodiment of the invention each of the transconductors may comprise a linearised differential amplifier comprising at least two emitter-coupled transistors with a bias-current source, the bases of the transistors constituting the first input and the second input and the collectors constituting the first output and the second output. The first output and the second output of each transconductor may then be loaded by a current source. In the delay circuits described in the foregoing one of the inputs of the transconductors of the all-pass network is connected to earth and the signal current is taken from both outputs of the transconductors. An analysis of these circuits has revealed that the circuits operate in the same way if both inputs of the transconductor are floating and signal current is taken from only one of the outputs. In that case a delay circuit comprising at least one all-pass network having an input terminal for receiving an input signal, an output terminal for supplying an output signal, and a reference terminal for carrying a reference potential, and comprising an at least second-order filter section which comprises three transconductors, which each have a first input, a second input, a first output and a second output, and a first and a second capacitor, may be characterized in that the first output of the first transconductor is coupled to the input terminal, the first input of the second transconductor is coupled to the first output of the first transconductor, the first output of the second transconductor is coupled both to the second input of the first transconductor and to the first input of the third transconductor, and the second input of the second transconductor is coupled to the first output of the third transconductor, in that the first capacitor is arranged between the first output of the first transconductor and that of the second transconductor and the second capacitor is arranged between the first output of the first transconductor and that of the third transconductor, and in that a third capacitor is arranged between the first output of the second transconductor and that of the third transconductor. In this embodiment the presence of the third capacitor also reduces the likelihood that resonance effects occur. When the delay circuit is thus constructed each all-pass network may comprise a first-order filter section which together with the second-order filter section constitutes a third-order filter section, which first-order filter section comprises a fourth transconductor having a first input, a second input, a first output and a second output, and a fourth capacitor. In this delay circuit the likelihood that resonance effects occur may also be further reduced if the delay circuit is characterized in that the fourth transconductor has its first input coupled to the first output of the third transconductor and its first output to the second input of the third transconductor, in that the fourth capacitor is arranged between the first output of the third transconductor and that of the fourth transconductor, and in that the third-order section comprises a fifth capacitor arranged between the first output of the second transconductor and that of the fourth transconductor, and a sixth capacitor which is arranged between the first output of the first transconductor and that of the fourth transconductor. In yet another embodiment each all-pass network may comprise a cascade of at least two second-order filter sections, the third transconductor of the first section and the first transconductor of the second section being formed by a single transconductor which is common to both sections. If the delay circuit comprises a plurality of cascaded networks, in accordance with a further embodiment, the transconductor of a network which is connected to the output terminal and the transconductor of a subsequent network which connected to the input terminal may be formed by a single transconductor which is common to both sections. In such delay circuits, in a further embodiment, each of the transconductors may comprise a linearised differential amplifier comprising at least two emitter-coupled transistors with a bias-current source, the bases of the transistors constituting the first input and the second input and the collectors constituting the first output and the second output. Said first output and second output of each transconductor may be coupled to each other by means of a current mirror or a negative immittance converter. Embodiments of the invention will now be described in more detail, by way of example, with reference to the accompanying drawings, in which Fig. 1a is a basic diagram of a delay line comprising a delay circuit, Fig. 1b a basic diagram of a transversal filter comprising a delay circuit, Fig. 2a shows a delay circuit in accordance with a first embodiment of the invention, comprising a second-order all-pass network, Fig. 2b is a diagram to explain the structure of the circuit shown in Fig. 2a, Fig. 3a shows a delay circuit in accordance with the invention comprising two cascaded second-order networks as shown in Fig. 2a, Fig. 3b is a diagram to explain the structure of the circuit shown in Fig. 3a, Fig., 4 shows a delay circuit in accordance with the invention comprising a first type of third-order all-pass network, Fig. 5 shows a delay circuit in accordance with the invention comprising an improved third-order all-pass network of the first type, Fig. 6 shows some transconductors for use in delay circuits in accordance with the invention, Fig. 7 shows the circuit of Fig. 5 comprising transconductors as shown in Fig. 6, Figs. 8a, b and c show some diagrams to analyzes the operation of the circuit shown in Fig. 2a, Fig. 8d shows a delay circuit in accordance with the invention comprising an all-pass network of a second type, Fig. 9a shows a delay circuit in accordance with the invention comprising two cascaded second-order networks as shown in Fig. 8d, Fig. 9b shows a delay circuit in accordance with the invention comprising a third-order all-pass network of a second type, Fig. 9c shows a delay circuit in accordance with the invention comprising an improved third-order network of the second type, Fig. 10a shows a transconductor of a first type for use in the delay circuits of Fig. 8d and Fig. 9, Fig. 10b shows a transconductor of a second type for use in the delay circuit shown in Fig. 8d and Fig. 9, Fig. 11a shows a first elaborated version of the circuit shown in Fig. 10b, Fig. 11b shows a second elaborated version of the circuit shown in Fig. 10b and Fig. 12 shows the circuit of Fig. 9c comprising transconductors as shown in Fig. 11a. Fig. 1a is the basic diagram of a delay line for which the delay circuit in accordance with the invention can be used. The delay line comprises a plurality of all-pass networks D₁, ..., D n arranged in cascade. In this respect an all-pass network is to be understood to mean a network whose gain is substantially constant and equal to unity over a specific bandwidth. Each network D k (k = 1, ..., n) has an input terminal a k , an output terminal b k and a reference terminal r k . The reference terminal r k of each network is connected to an earth terminal 3. The input terminal a k of each network D k is connected to the output terminal b k-1 of the preceding network D k-1 , except for the input terminal a₁ of the first network D₁, which constitutes the input terminal of the delay line and which receives the input voltage V i . The delayed output voltage V o appears across a load resistor R L arranged between the output terminal b n of the last network D n and the earth terminal 3. The delay time of each of the networks D k may be fixed, but it is also possible to make the delay time variable by connecting a control input of each of the networks to a control circuit 4. Fig. 1b shows the basic diagram of a transversal filter in which the delay circuit in accordance with the invention may be employed. Identical parts bear the same reference numerals as in Fig. 1a. In such a filter the input signal V i and the signals delayed by the networks D k are multiplied by a weighting factor in weighting means W k (k = 0, 1, ..., n) and are subsequently added to each other in a summing device 5. On the output of this device this yields an output signal which is a filtered replica of the input signal V i . An adaptive filter can be obtained by making the weighting factors variable. Fig. 2a shows an example of an all-pass network for a delay circuit in accordance with the invention, which network comprises a second-order filter section. For the sake of simplicity the Figure only shows the signal paths of the circuit and bias-current sources and the like are not shown. The second-order filter section comprises three identical transconductors G₁, G₂ and G₃, which are shown in their simplest form, namely a different amplifier comprising two emitter-coupled transistors, whose bases constitute the inputs and whose collectors constitute the outputs of the transconductor. The first transconductor G₁ comprises the transistors T₁ and T₂, the base of the transistor T₁ constituting the first input 10, which is connected to the earh terminal 3, and the base of the transistor T₂ constituting the second input 11, which is connected to the input terminal a₁ of the network to which the input signal is applied. The collector of the transistor T₁ constitutes the first output 12 and the collector of the transistor T₂ constitutes the second output 13 of the transconductor G₁. The second transconductor G₂ comprises the transistors T₃ and T₄, the transistor T₃ having its base 14 connected to the earth terminal 3 and its collector 16 to the base of the transistor T₂ and the transistor T₄ having its base 15 connected to the collector 13 of the transistor T₂. The third transconductor G₃ comprises the transistors T₅ and T₆, the transistor T₅ having its base 18 connected to the earth terminal 3 and its collector 20 to the base of the transistor T₄ and the transistor T₆ having its base 19 connected to the collector 17 of the transistor T₄ and to the output terminal b₁ of the network. Further, the all-pass network comprises a first capacitor C₁ arranged between the bases 11 and 15 of the transistors T₂ and T₄, and a second capacitor C₃ arranged between the bases 11 and 19 of the transistors T₂ and T₆. The network further comprises a third capacitor C₂ which is arranged between the bases 15 and 19 of the transistors T₄ and T₆. The addition of the capacitor C₂ to the second-order network reduces resonance effects in the network because resonant-rise symptoms caused by parasitic effects, for example as a result of stray capacitances, are reduced. These parasitic effects inter alia produce a rise in gain, so that the amplitude characteristic of the network is not as flat as possible and is not substantially equal to unity within the desired frequency range. A voltage VI is applied to the input terminal a₁ in order to determine the transfer characteristic of the network. The collector 12 of the transistor T₁ is then connected to the earth terminal 3. Further, the network is loaded by a load resistance, for which purpose the collector 21 of the transistor T₆ is connected to its base 19. By means of Fig. 2b it will be demonstrated that by making this connection the network is loaded by a resistance. This Figure only shows the last transconductor of the network. The collector of the transistor T₆ is connected to the earth terminal 3. A resistance R between the output terminal b₁ and the earth terminal 3 is constituted by a transconductor which is identical to the transconductor G₃ and which comprises the transistors T₇ and T₈ whose bases are connected to their collectors. Since the bases of the transistors T₆ and T₈ are at the same voltage, the current through the resistance R will be the same as that through the transconductor G₃. The current through the resistance R is drained to the earth terminal 3, because the collector of the transistor T₇ is connected to the earth terminal 3. The transistor T₆ only supplies current to the transistor T₅, because the collector of the transistor T₆ is connected to the earth terminal 3. Since the collector currents of the transistors T₆ and T₇ do not serve any useful purpose, these transistors may be dispensed with. The emitter of the transistor T₈ should then be connected to the emitter of the transistor T₅. This results in the configuration shown in Fig. 2a. When the voltage VI is applied an input current IO will flow through the input terminal a₁, which current is entirely drained to earth via the collector 12 of the transistor T₁. This means that this current is equal to IO = GVI, where G is the transconductance of the transconductor G₁. Consequently, the input impedance of the network is equal to 1/G. As this is a pure resistance, the networks may be coupled directly to one another. If the voltages on the inputs 15 and 19 are V2 and V3, respectively and the base currents of the transistors T₄ and T₆ are ignored, the sum of the currents through the inputs 15 and 19 is equal to: -GV1 + GV3 + (V1 - V2)pC1 + (V3 - V2)pC2 = 0 (1) -GV2 - GV3 + (V2 - V3)pC2 + (V1 - V3)pC3 = 0 (2) where p = o + jw is the complex frequency, in which o is the attenuation and w is the angular frequency. From the above equations it follows that the transfer function of the all-pass network is: From this equation it is easy to derive what the gain of the network is unity for all frequencies. The phase shift of the network as a function of the frequency can also be derived from the above equation, which enables the delay function, i.e. the delay time as a function of the frequency, to be computed. For a second-order delay network the polynomials in p in the numerator and the denominator of equation (3) are of the second degree, enabling a resonant frequency W o and a quality factor Q to be defined. As is known, the transfer function of a second-order all-pass network, expressed in these quentities, may be written as: It follows from equations (3) and (4) that: In a network comprising one second-order section the resonant frequency W o and the quality factor Q may be selected, for example, in such a way that the delay function is as flat as possible or that the delay time for all the frequencies within the bandwidth of the signal to be delayed equals the desired delay time as closely as possible. When a chain of such networks is integrated it is important to minimise the overall capacitance of a network in view of the required chip area. From the equation for the resonant frequency it follows that for a given transconductance G the sum of the capacitances C1 + C2 is defined by the selected resonant frequency W o and the quality factor Q. From the equation for the quality factor Q it follows that the capacitance C3 is minimal if C1 = C2. when it is assumed that C1 = C2 = C/2, it follows from the foregoing that As a result of the above choice the overall capacitance of a network is minimised, whilst at the same time it is found that resonance effects are minimised owing to the choice C1 = C2. Fig. 3a shows a delay circuit comprising two second-order all-pass networks, in which identical parts bear the same reference numerals as in Fig. 2. It is to be noted that in reality the filter circuit may comprise several tens of networks. The second network D₂ is constructed in the same way as the network D₁ and comprises three transconductors G₃, G₄ and G₅. The transconductor G₃ is a transconductor which is common to the network D₁. This will be explained with reference to Fig. 3b, in which if the first network D₁ and the second network D₂ each comprise three transconductors the last transconductor G₃ is shown to comprise transistors T₅ and T₆ of the network D₁ and the first transconductor G 3a is shown to comprise the transistors T 5a and T 6a . The collectors of the transistors T₆ and T 5a are connected to the earth terminal 3. Therefore, the collector currents of these transistors are not used for any further purpose in this circuit. These collector currents are equal, because the two transconductors have equal transconductances and the voltage across the two transconductors is the same. Therefore, the operation of the circuit does not change if the transistor T 6a supplies current to the transistor T₅ and the transistors T₆ and T 5a are dispensed with. The last transconductor G₅ of the network D₂ is loaded by a resistance in the same way as the transconductor G₃ in Fig. 2a. The circuit of Fig. 3a is driven by a signal-current source I across which a source resistance R o is connected. This resistance is constituted by a transconductor G o whose outputs are connected to its inputs. In order to minimise the effect of, for example, tolerances the transconductance of this transconductor G o is suitably the same as that of the transconductors in the networks D₁ and D₂. It is to be noted that the delay circuit may also be driven by a signal-voltage source, in which case the resistance R o should be connected in series with the voltage source. For example, in the above manner a delay line has been realised to delay video signals with a bandwidth of 0-5 MHz and a delay of 0.1 µs for each all-pass network, the maximum deviation from the desired delay being 9.7%. The resonant frequency and the quality factor of each network were equal to W o = 4.98 MHz and Q = 0.702 respectively, the capacitance of the capacitors C1, C2 and C3 being equal to C1 = C2 = 22.762 pF and C3 = 11.054 pF for a transconductance G = 1 kOhm⁻¹ of the transconductors of the network. In the circuit shown in Fig. 3a each all-pass network comprises one second-order filter section, such a section in the present example being designed to provide a delay which is as constant as possible over the desired bandwidth. However, it is also possible to construct each all-pass network from a plurality of, for example, two, second-order filter sections which within a specific frequency range each have a delay with a comparatively strong frequency-dependence and which in combination provide a substantially constant delay over the desired bandwidth. For example, in this way a delay line has been realised in which the networks were each constructed from two cascaded second-order filter sections. Fig. 3a shows one all-pass network for this case. The networks had a delay of 0.2 µs over a bandwidth of 0-5 MHz, the maximum deviation being 0.73%. The first second-order filter section, reckoned from the input of the delay line, had a resonant frequency f₁ = 3.72 MHz and a quality factor Q₁ = 0.569 and the capacitances of the capacitors were equal to C1 = C2 = 37.595 pF and C3 = 5.546 pF, whilst the second-order filter section had a resonant frequency f₂ = 5.81 MHz and a quality factor Q₂ = 1.129 and the capacitors had capacitances equal to C1 = C2 = 12.132 pF and C3 = 24.861 pF. The transconductance of the transconductors of both the first section and the second section was G = 1 kOhm⁻¹. Fig. 4 shows an all-pass network for a delay circuit in accordance with the invention comprising a third-order filter section. This third-order filter section comprises a first-order filter section D₁ and a second-order filter section D₂. The first-order section D₁ comprises a transconductor G₁₀ comprising the transistors T₁₀ and T₁₁ and a capacitor C₁, and a transconductor G₁₁ which it has in common with the second-order section D₂. The second-order section D₂ is identical to that shown in Fig. 2 and comprises three transconductors G₁₁ comprising transistors T₁₂, T₁₃, G₁₂ comprising the transistors T₁₄, T₁₅ and G₁₃ comprising the transistors T₁₆, T₁₇, and three capacitors C₂, C₃ and C₅. By means of such a third-order all-pass network it is possible to obtain a delay which is equal to the desired delay over the entire bandwidth with an accuracy higher than in the case of a second-order all-pass network. For example, networks have been realised which exhibit a delay of 0.1 µs over a bandwidth of 0-5 MHz with a maximum deviation of 0.22%. The pole frequency of the first-order section was f₁ = 6.40 MHz and the first capacitor had a capacitance of C₁ = 24.868 pF, whilst the resonant frequency and the quality factor of the second-order section were f₂ = 8.10 MHz and Q₂ = 0.779 respectively, the capacitances of the capacitors being C2 = C3 = 12.611 pF and C5 = 9.001 pF. The transconductors G₁₀ to G₁₃ inclusive again had transconductances equal to G = 1 kOhm⁻¹. In this third-order all-pass network the tendency to resonate as a result of stray capacitances can be reduced further by means of two additional capacitors, as is shown in Fig. 5. In this Figure a capacitor C4 is arranged between the inputs 25 and 27 of the transconductors G₁₀ and G₁₂ and a capacitor C6 between the inputs 25 and 28 of the transconductors G₁₀ and G₁₃. It is to be noted that said capacitor C6 may be dispensed with. The transfer function of this network can be derived by applying an input voltage V₁, which, in the same way as in the network shown in Fig. 3a, results in an input current IO which is drained to the earth terminal 3 via the first output of the first transconductor G₁₀. If the voltages on the inputs 26, 27 and 28 are V2, V3 and V4 and the base currents of the transistors of the transconductors are ignored, the following equations may be drawn up for the currents through the inputs 26, 27 and 28: -GV1 + GV3 + (V1 - V2)pC1 + (V3 - V2)pC2 + (V4-V2)pC5 = 0 (7) -GV2 + GV4 + (V1 - V3)pC4 + (V2 - V3)pC2 + (V4-V3)pC3 = 0 (8) -GV3 - GV4 + (V1 - V4)pC6 + (V2 - V4)pC5 + (V3-V4)pC3 = 0 (9) From the above equations it follows that the transfer function of the all-pass network is: where A = C1 + C2 + C3 + C6 B = C1.C2+C1.C3+C1.C4+C2.C3+C2.C4+C2.C5+C3.C5+C4.C5 D = C1.C2.C3 + C1.C2.C5 + C1.C3.C4 + C1.C3.C5 + C1.C4.C5 + C2.C3.C4 + C2.C4.C5 + C3.C4.C5 + BC6. From this equation it is evident that the gain of the network is equal to unity for all frequencies. In view of integration it is again important minimise the overall capacitance of the network. Minising the overall capacitance can now be effected only numerically. In this way the following capacitance values have been found for a delay line for a video signal having a bandwidth of 0-5 MHz and a delay of 0.1 µs per network for a transconductance G = 1 kOhm⁻¹ of the transconductors: C1 = C2 = C3 = 16.246 pF C4 = C5 = 3.247 pF C6 = 1.354 pF. In the above examples the transconductors are always represented as simple differential amplifiers. In order to enable comparatively high input voltages to be handled the transconductance should be contant over a maximum voltage range. In a simple differential amplifier the output current increases as a linear function of the input voltage over a small range only, so that the transconductance is only constant over a small range of input voltages. Therefore, linearised differential amplifiers are generally used as transconductors. An example of such a transconductor shown in Fig. 6a. The transconductor comprises two differential amplifiers arranged in parallel. The first differential amplifier comprises the transistors T₂₀ and T₂₁, which transistors have unequal emitter areas. A current source I₁ = I is arranged in the common emitter line of these transistors. The second differential amplifier comprises the transistors T₂₂ and T₂₃, which have also unequal emitter areas. The ratio between the emitter areas of the transistors T₂₀ and T₂₁ is equal to that of the transistors T₂₃ and T₂₂. In the common emitter line of the transistors T₂₂ and T₂₃ a current source I₂ = I is arranged. The commoned bases of the transistors T₂₀ and T₂₂ constitute a first input 30 and the commoned bases of the transistors T₂₁, T₂₃, constitute a second input 31 of the transconductor. The collectors of the transistors T₂₀ and T₂₂ are connected to a first output 32 and the collectors of the transistors T₂₁ and T₂₃ are connected to a second output 33. The outputs 32 and 33 are loaded by a current source I₃ = I and I₄ = I respectively. If the ratio between the emitter areas of the transistors T₂₀ and T₂₁ is 4 the transconductance, which is equal to the ratio between the signal current on the outputs 32 and 33 and the signal voltage across the inputs 30 and 31, is given by G = 8 25 qI kT . The linear range of such a differential amplifier is approximately 5 times as large as that of a single differential amplifier. From the equation for the transconductance it also follows that this transconductance can be varied by varying the quiescent current I. The delay of the network can be varied by varying this current for all the transconductors in a delay circuit. Fig. 6b shows a second example of a linearised transconductor. It comprises a first differential amplifier comprising the transistors T40 and T41, in whose common emitter line a current source I₁ = 2I is arranged, and a second differential amplifier comprising the transistors T₄₂ and T₄₃ in whose common emitter line a current source I₂ = 2I is arranged. The bases of the transistors T₄₀ and T₄₂ are connected to an input 40, to which the input voltage V in is applied. The base of the transistor T₄₁ is connected to a terminal 41 to which a reference voltage V R1 is applied and the base of the transistor T₄₃ is connected to a terminal 42 to which a reference voltage V R2 is applied. The collectors of the transistors T₄₀, T₄₂ are connected to a first output 43, which is loaded by a current source I₃ = (I₁ + I₂)/2 and the collectors of the transistors T₄₁, T₄₃ are connected to a second input 44, which is loaded by a current source I₄ = (I₁ + I₂)/2 the reference voltages V R1 and V R2 are symmetrical about a voltage level V R0 of the filter-earth. By a suitable choice of these voltages it is possible to achieve that the output current increases as a linear function of the input voltage over a wider range than attainable with a single differential amplifier. This linear range can be extended further if, as is illustrated in Fig. 6c, a third differential amplifier comprising the transistors T44 and T45 is added to the circuit, a current source I₃ = k.2I (k < 1) being arranged in the common emitter line of these transistors. The base and the collector of the transistor T44 are connected to the input 40 and to the output 43, respectively. The base of the transistor T45 is connected to a terminal 45, to which a reference voltage V R0 is applied. The collector of the transistor T45 is connected to the output 44. In the case of a suitable choice of the voltages V R1 and V R2 , which are again symmetrical about the value V R0 , and of the ratio between the currents of the current sources I₁, I₂ and I₃, a comparatively wide range of constant transconductance is obtained. The linear range of the transconductor can be further extended by adding further differntial amplifiers to the circuit. In Fig. 7 the network of Fig. 5 comprises transconductors of the type shown in Fig. 6. It is to be noted that the invention is not limited to the transconductor shown but that it is alternatively possible to utilize other transconductors such as those described in the previously filed Patent Application (PHN 11.660). Fig. 8a again shows the circuit of Fig. 2a, the capacitors being omitted for the sake of clarity. The current I2 applied to the junction point 15 is equal to the difference between the collector currents of the transistors T₂ and T₅. This means that this current complies with: I2 = GV1 - GV3 = G(V1 - V3) (11) An analysis of this equation reveals that the same current I2 can also be obtained by applying a voltage V3 - V1 across a transconductor having a transconductance G, as is indicate n Fig. 8b. In the present case the transconductor does not comprise an earthed input as in the case of the transconductor shown in Fig. 8a and further it comprises one instead of two outputs from which the signal current can taken. The other output is connected to earth. The following equations are valid for the currents through the junction points 11 and 19: I1 = -GV2 = G(0 - V2) I3 = GV2 + GV3 = G(V2 - 0) + G(V3 - o) (12) Thus, these currents can be obtained by applying the relevant voltages across transconductors. The circuit of Fig. 8a modified in accordance with this principle is shown in Fig. 8c, G L being the transconductor constituting the load resistance. Fig. 8b shows the complete second-order all-pass network. Further this circuit operates in the same way as that shown in Fig. 2. Figs. 9a, b and c show the circuits of Figs. 3, 4 and 5 in similarly modified form. It is to be noted that also for this type of transconductors cascading of identical networks enables one common transconductor to be obtained, so that one transconductor can be saved. In the circuits shown in Figs. 8 and 9 signal current is taken from only one output of the transconductor. The other output is not used and the signal current which is in principle available on this output is drained to a supply-voltage terminal. However, the output signal current of the transconductor may be increased by adding the signal current from the non-used output to the output signal current. In principle, this can be achieved by means of a current mirror. Fig. 10a shows the transconductor of Fig. 6 provided with a current mirror comprising a diode-connected PNP transistor T₃₀ and a PNP transistor T₃₁. However, the use of PNP transistors is less desirable because they have an adverse effect on the high-frequency behaviour of the circuit. Fig. 10b illustrates a solution which is more favourable in this respect. Here, the transconductor is loaded by a negative immittance converter comprising an operational amplifier 36 having an inverting input (-) connected to the junction point 33, a non-inverting input (+) connected to the junction point 32, and an output 34 connected to the inverting input via a resistor R2 = R and to the non-inverting input via a resistor R1 = R. The junction point 32 is connected to the output 35 of the transconductor. The signal currents through the junction point 33 and the junction point 32 are in phase opposition. The immittance converter converts the signal current through the junction point 33 and adds this current to the signal current through the junction point 32, so that the output signal current is doubled. This is because the operational amplifier 30 ensures that the voltage on the inverting input is equal to that on the non-inverting input. As the resistors R₁ and R₂ are equal, the currents through these resistors are equal. The current through the junction point 33 flows through the resistor R₂, so that the same current also flows through the resistor R₁. Fig. 11 shows two examples of such a transconductor comprising a negative immitance converter. In Fig. 11a the junction point 33 between the collectors T₂₁ and T₂₃ is connected to the positve power-supply terminal by means of the series arrangement of the resistor R₂, a diode-connected transistor T₃₃, and a current source I₅ = I + J, and the junction point 32 between the collectors of the transistors T₂₀ and T₂₂ is connected to the positive power-supply terminal by means of the series arrangement of the resistor R₁ and the collector-emitter path of a transistor T₃₂ having its base connected to the base of the transistor T₃₃. Further, the junction points 32 and 33 are connected to the bases of two transistors T₃₄ and T₃₅ arranged as a differential amplifier, whose common emitter terminal is connected to the negative power-supply terminal via a current source I₆ = 2J. The collector of the transistor T₃₅ is connected to the base of the transistor T₃₂ and the collector of the transistor T₃₄ is connected to the positive power-supply terminal. Instead of the resistors R₁ and R₂ and the transistors T₃₂ and T₃₃ the circuit shown in Fig. 11b comprises two diodes D₁ and D₂ arranged respectively between the output and the non-inverting input 32 and between the output and the inverting input 33 of the differential amplifier T₃₄, T₃₅. The operation of the circuits shown in Figs. 11a and b is the same as that of the circuit shown in Fig. 10b. Finally, Fig. 12 shows the detailed circuit diagram of a third-order all-pass network as shown in Fig. 9c provided with transconductors as shown in Fig. 11a. Here, the outputs of the transconductors G₁₀ to G₁₃ and G L are moreover clamped at a voltage of 0 V by means of a clamping circuit comprising a NPN transistor T₄₀ whose emitter is connected to the output 33 of the relevant transconductor, whose collector is connected to the positive power-supply terminal, and whose base is connected to the terminal carrying the positive reference voltage V P , and a PNP transistor T₄₁ having its emitter connected to the output 33, having its collector connected to the negative power-supply terminal, and having its base connected to a terminal carrying a negative reference voltage V N . The reference voltages are, for example, V p = 2 D/3 , V N = -2 D/3, D being a diode voltage. When the requirement is met that the current J from the current suorce I₆ of the negative immittance converter is larger than the current I from the currents sources I₁ and I₂ of the transconductor, this voltage clamping ensures that after switching on of the power supply the circuit operates in the desired manner and that "latch-up" is not possible.

Claims: A delay circuit comprising at least one all-pass network which has an input terminal (a₁) for receiving an input signal, an output terminal (b₁) for supplying an output signal, and a reference terminal (3) for carrying a reference potential and which comprises an at least second-order filter section comprising three transconductors (G₁, G₂, G₃), which each have a first input (10, 14, 18), a second input (11, 15, 19), a first output (12, 16, 20) and a second output (13, 17, 21), the first input (10, 14, 18) of each of the three transconductors being coupled to the reference terminal (3), the second input (11) of the first transconductor (G₁) being coupled to the input terminal (a₁), the second input (15) of the second transconductor (G₂) being coupled to the second output (13) of the first transconductor (G₁) and to the first output (20) of the third transconductor (G₃), the second input (19) of the third transconductor (G₃) being coupled to the second output (17) of the second transconductor (G₂) and to the output terminal (b₁), which filter section further comprises a first capacitor (C₁) arranged between the second inputs (11, 15) of the first transconductor (G₁) and the second transconductor (G₂), and a second capacitor (C₃) arranged between the second inputs (11, 19) of the first transconductor (G₁) and the third transconductor (G₃), characterized in that the filter section comprises a third capacitor (C₂) which is arranged between the second inputs (15, 19) of the second transconductor (G₂) and the third transconductor (G₃) (fig 2a). A delay circuit as claimed in Claim 1, characterized in that the all-pass network further comprises a first-order filter section which together with the second-order filter section constitutes a third-order filter section, which first-order filter section comprises a fourth transconductor (G₁₃) having a first input, a second input (28), a first output (27) and a second output, and a fourth capacitor (C₃) (fig 4). A delay circuit as claimed in Claim 2, characterized in that the fourth transconductor (G₁₃) has its first input coupled to the reference terminal (3), its second input (28) to the second output of the third transconductor (G₁₂), its first output to the second input (27) of the third (G₁₂) transconductor, and its second output to the output terminal of the network, in that the fourth capacitor (C₃) is arranged between the second inputs (27, 28) of the third transconductor (G₁₂) and the fourth transconductor (G₁₃), and in that the third-order filter section further comprises a fifth (C₅) capacitor which is arranged between the second inputs (26, 28) of the second transconductor (G₁₁) and the fourth transconductor (G₁₃) (fig 4). A delay circuit as claimed in Claim 3, characterized in that the third-order filter section further comprises a sixth capacitor (C₆) arranged between the second inputs (25, 28) of the first transconductor (G₁₀) and the fourth transconductor (G₁₃) (fig 5). A delay circuit as claimed in Claim 1, characterized in that each all-pass network comprises a cascade of at least two second-order filter sections, the third transconductor of the first section and the first transconductor of the second section being formed by a single transconductor (G₃) which is common to both sections (fig 3a). A delay circuit as claimed in any one of the preceding Claims, characterized in that the delay circuit comprises a plurality of cascaded all-pass networks, every transconductor of an all-pass network which is connected to an output terminal and the transconductor of a subsequent all-pass network which is connected to the input terminal being formed by a single transconductor which is common to both sections. A delay circuit as claimed in any one of the preceding Claims, characterized in that the first output (12) of the transconductor (G₁) of the first all-pass network which is connected to the input terminal (a₁) is connected to the reference terminal (3) (fig 2a). A delay circuit as claimed in any one of the preceding Claims, characterized in that an input resistance is arranged between the input terminal of a first all-pass network of the delay circuit and the reference terminal (3), which input resistance is constituted by a fifth transconductor (G₀) having a first input, a second input, a first output and a second output, the first input being connected to the first output and the second input being connected to the second output (fig 3a). A delay circuit as claimed in any one of the preceding Claims, characterized in that the output terminal of the last all-pass network of the delay circuit is connected to the second output of the transconductor (G₅) connected to the output terminal (fig 3a). A delay circuit as claimed in any one of the preceding Claims, characterized in that each of the transconductors comprises a linearised differential amplifier comprising at least two emitter-coupled transistors (T₂₀, T₂₁) with a bias-current source (I₁), the bases (30, 31) of the transistors constituting the first input and the second input and the collectors (32, 33) constituting the first output and the second output (fig 6a, fig 7). A delay circuit as claimed in Claim 10, characterized in that the first output and the second output of each transconductor are loaded by a current source (I₃, I₄) (fig 6a, 7). A delay circuit comprising at least one all-pass network having an input terminal (V₁) for receiving an input signal, an output terminal (V₃) for supplying an output signal, and a reference terminal for carrying a reference potential, and comprising an at least second-order filter section which comprises three transconductors (G₁, G₂, G₃), which each have a first input, a second input, a first output and a second output, and a first (C₁) and a second (C₃) capacitor, characterized in that the first output of the first transconductor (G₁) is coupled to the input terminal (V₁), the first input of the second transconductor (G₂) is coupled to the first output (V₁) of the first transconductor (G₁), the first output (V₂) of the second transconductor (G₂) is coupled both to the second input of the first transconductor (G₁) and to the first input of the third transconductor (G₃), and the second input of the second transconductor (G₂) is coupled to the first output (V₃) of the third transconductor (G₃), in that the first capacitor (C₁) is arranged between the first output (V₁) of the first transconductor (G₁) and that (V₂) of the second transconductor (G₂) and the second capacitor (C₃) is arranged between the first output (V₁) of the first transconductor (G₁) and that (V₃) of the third transconductor (G₃), and in that a third (C₂) capacitor is arranged between the first output (V₂) of the second transconductor (G₂) and that (V₃) of the third transconductor (G₃) (fig 8d). A delay circuit as claimed in Claim 12, characterized in that each all-pass network comprises a first-order filter section which together with the second-order filter section constitutes a third-order filter section, which first-order filter section comprises a fourth transconductor (G₁₃) having a first input, a second input, a first output and a second output, and a fourth capacitor (C₃) (fig 9b). A delay circuit as claimed in Claim 13, characterized in that the fourth transconductor (G₁₃) has its first input coupled to the first output of the third transconductor (G₁₂) and its first output to the second input of the third transconductor (G₁₂), in that the fourth capacitor (C₃) is arranged between the first output of the third (G₁₂) transconductor and that of the fourth (G₁₃) transconductor, and in that the third-order section comprises a fifth (C₅) capacitor arranged between the first output of the second (G₁₁) transconductor and that of the fourth (G₁₃) transconductor (fig 9b). A delay circuit as claimed in Claim 14, characterized in that the third-order section further comprises a sixth (C₆) capacitor arranged between the first output of the first transconductor (G₁₀) and that of the fourth transconductor (G₁₃) (fig 9c). A delay circuit as claimed in Claim 12, characterized in that each all-pass network comprises a cascade of at least two second-order filter sections the third transconductor of the first section and the first transconductor of the second section being formed by a single transconductor (G₃) which is common to both sections (fig 9a). A delay circuit as claimed in any one of the preceding Claims 12 to 16, characterized in that the delay circuit comprises a plurality of cascaded all-pass networks, the transconductor of a network which is connected to the output terminal and the transconductor of a subsequent network which is connected to the input terminal being formed by a single transconductor which is common to both sections. A delay circuit as claimed in any one of the preceding Claims 12 to 17, characterized in that the first output of the transconductor (G₁) of the first all-pass network which is connected to the input terminal is connected to the first input of said transconductor (fig 9a). A delay circuit as claimed in any one of the preceding Claims 12 to 18, characterized in that a load (G L ) is arranged between the output terminal of the last all-pass network of the delay circuit and the reference terminal, which load is constituted by a fifth transconductor (G L ) having first input, a second input, a first output and a second output, the first input and the first output being connected to the output terminal and the second input being connected to the reference terminal (fig 9a). A delay circuit as claimed in any one of the preceding Claims 12 to 19, characterized in that each of the transconductors comprises a linearised differential amplifier comprising at least two emitter-coupled transistors (T₂₀, T₂₁) with a bias current source (I₁), the bases of the transistors constituting the first input (30) and the second input (31) and the collectors constituting the first output (32) and the second (33) output (fig 10a). A delay circuit as claimed in Claim 20, characterized in that the first output and the second output of each transconductor are coupled to each other by means of a current mirror (T₃₀, T₃₁) (fig 10a). A delay circuit as claimed in Claim 20, characterized in that the first output (32) and the second (33) output of each transconductor are coupled to each other by means of a negative immittance converter (36, R₁, R₂) (fig 10b). A delay circuit as claimed in Claim 10 or 20, characterized in that each of the transconductors comprises two differential amplifiers which are arranged in parallel and which each comprise two transistors having unequal emitter areas, the bases and the collectors of each time two transistors having unequal emitter areas being interconnected. A delay circuit as claimed in Claim 10 or 20, characterized in that each of the transconductors comprises at least two differential amplifiers which each comprise two transistors having substantially equal emitter areas, the bases and the collectors of one of the transistors of each differential amplifier being interconnected, the bases of the other transistors of each differential amplifier carrying mutually different reference voltages, and the collectors of said other transistors being interconnected.

Inventors: SNIJDER, PIETER JACOB, VOORMAN, JOHANNES OTTO, VROMANS, JOHANNES SOPHIA
Applicants: N.V. PHILIPS' GLOEILAMPENFABRIEKEN
Application year: 1987
Main classification: H03H11
IPC codes: H04B3, H03H11, H03H15