DC TO DC CONVERTER CURRENT PUMP

Abstract: A DC to DC power converter designated a synchronous current pump (13) and operated in the preferred mode in synchronization with a discharge circuit (11) and using a capacitor (28) as the energy storage element, and in the preferred embodiment has in series with said capacitor the battery supply (10), an inductor (30), a diode (27a), and the primary winding (31a) of a transformer (31); and across said storage capacitor is an energy transfer FET switch (33) which is used for discharging said capacitor and transferring its stored energy to an output load capacitor (4) connected through a diode (32) to the secondary winding of said transformer. In operation, the current pump supplies power efficiently and smoothly to a load discharge capacitor in synchronization with operation of the discharge circuit.

Description: DC TO DC CONVERTER CURRENT PUMP FIELD OF THE INVENTION The present invention relates to DC to DC converter power supply systems. BACKGROUND OF THE INVENTION AND PRIOR ART The need for, and actual designs of, DC to DC converters for converting automobile battery voltage to a higher voltage for charging a capacitor of a capacitive discharge ignition system are known in the prior art. These converters generally operate on the principle of switching an inductive current and are thus sensitive to load conditions, are generally not as efficient as desired, and are often more complex than desired. The provision of "boost power" during the ignition firing period (which consists of several ignition pulses separated by non-pulsing periods) so as to reduce the decay rate of the voltage of the various pulses is disclosed herein by me as a base¬ line for further improvements disclosed herein. The present invention comprises a novel form of DC to DC converter power supply which, among other things, uses a capacitor and not an inductor as the storage element, and is load insensi¬ tive, is efficient, is simple, and is to able to operate synchro¬ nously with the operation of the ignition discharge circuit and thus provide some level of power boost even at moderate power levels. -2- OBJECTS OF THE INVENTION It is a principal object of this invention to provide a power supply with a capacitor as the energy storage element, and which is simple and practical and provides power smoothly to an output load capacitor which may be discharged in a time regular way, the power supply called a "synchronous current pump" when operated in synchronization with said time regular discharge; said current pump made up of a large capacitor, a transformer, an inductor, diodes and switches, and operating as a CD circuit providing "slugs" of current synchronously with the output CD circuit firing. In general, because of the power supply's gene¬ rally synchronous operation with the load capacitor discharge, it in effect also provides "boost power" or high power * during the spark pulsing period to maintain a higher voltage during this period. Another object is to provide several forms of such current pump including a particular simple form (the simplified current pump) which has only one switch in its input circuit (i.e. the battery-side switch is replaced with a diode and an inductor, if an inductor is not otherwise present). Another object is to provide an optimized simplified current pump which includes across the transformer secondary winding in parallel with the output load capacitor and diode a small tuning and feedback capacitor whose value is precisely determined by circuit principles to provide optimized current pump timing and positive electrical energy feedback, to provide high output power, very simple and low cost design, and a very high efficiency when the current pump is operated in a timed manner determined by this optimized circuit. Other features and advantages will be pointed out herein¬ after, and will become apparent from the following discussion including a Summary of the invention and Description of Particu¬ lar Preferred Embodiments of the invention when read in conjunc¬ tion with the accompanying drawings. -3- SUMMARY OF THE INVENTION This invention comprises a novel DC to DC converter (called a "current pump") used principally for charging a high voltage output capacitor of a capacitive discharge (CD) ignition circuit. The current pump uses a storage capacitor to store the energy, and itself operates on the principle of a CD circuit by using a shunt switch across the storage capacitor and the primary of a transformer to discharge the storage capacitor and charge the out¬ put or load capacitor connected to the output of the transformer. If used to provide power boost, then the current pump is prefer¬ ably operated synchronously with the ignition pulses to provide slugs of energy in between the pulse firings. The size of the current pump storage capacitor is much less in value to the the "transformed" output capacitor (generally the order of ten times greater than tne actual value of the output capacitor for a four hundred volt CD ignition system with an output capacitor in the range of 2 to 10 microFarads (uFarads)). Preferably FET semiconductor swithes are used for the main energy transfer (for tranferring energy from storage to load capacitor). In operation, the storage capacitor is charged from the battery to twice battery voltage (due to voltage doubling), and then discharged through a (FET) switch, the first half cycle of the energy discharge in general transferring most of the energy to the load capacitor. During the storage capacitor charging time or at any time excepting the first cycle of the energy discharge stage, the ignition capacitor may be discharging. In this way is provided a power supply for use especially with a CD system which is load insensitive, simple, efficient, and which can operate synchronously with the CD circuit for optimal transferring of energy to the load. As an optimized simplified current pump, this power supply represents a particularly simple power supply which uses positive feedback and precise tuning to provide a very high power output and very high efficiency. The invention may be.used in automobile and other ignition systems, and may be used in other than ignition systems of capa¬ citor load form. -4- BRIEF DESCRIPTION OF THE DRAWINGS The nature and objects of the invention are illustrated and described with reference to the following drawings, which also illustrate the preferred embodiments of the invention: FIG. 1 depicts a generic form of the current pump connected to a discharge circuit and including controls for regulating the output voltage and operating the discharge circuit and current pump. FIGS, la to Id depict preferred embodiments of the current pum . FIGS. 2a to 2e depict various waveforms corresponding tα the operation of both the simplified synchronous current pump and the discharge circuit during its pulsing period. A FIGS. 3a and 3b are circuit drawings of two forms of current pump DC-DC converter, a current pump with a switch in place of the input diode (FIG.3a), and a current pump including an extra SCR switch (FIG. 3b). FIG. 4 depicts a detailed circuit drawing of the synchronous current pump and the input trigger shaper/initiator necessary to produce both the initial ringing spark and to initiate the gate pulse width controller used also to provide "power boost". FIG. 5 depicts a circuit drawing of the optimized simplified current pump which is physically identical to that of FIG. la excepting for the absence of the output inductor in series with what is now defined as the tuning and feedback capacitor. FIGS. 5a to 5d depict various waveforms corresponding to the operation of the optimized simplified current pump shown in FIG.5. DESCRIPTION OF PREFERRED EMBODIMENTS FIG. 1 shows a generic form of the current pump type DC to DC power converter 13 designed to operate generally with fifty to one hundred watts output power (for a four stroke, four to eight cylinder engine) and with a high efficiency of about 80%. Preferably, the converter has the ability to provide higher peak or "boost" power during ignition, as is discussed below with reference to FIG. 3b and elsewhere. In its main application it is used for charging capacitor 4 of capacitance C (or Cl) micro- farads (uFarads) which is in series with primary winding 1 of a coil of a CD system 11 including SCR switching element 5 and diode 7, and the balance of which may include an ignition firing tip as shown in FIG. la or may comprise other loads. ... Current pump 13 operates once ignition switch 9 is closed by sequentially depositing energy provided by battery 10 into capa¬ citor 28 of capacitance Ce (or CO) through semiconductor element 26 and choke inductor 30, and discharging the stored energy through shunting switch 33 (FET shown) and transformer 31 (of turns ratio N) to charge capacitor 4 of CD circuit 11 connected to terminal 14a. Typically, the peak energy stored in capacitor 4 (C) is much greater than the peak stored in capacitor 28 (Ce), the parameter "Lambda" designating this ratio, which is given by: Lambda = ((N**2)*C)/Ce, where "**" means exponentiation. Typically Lambda is in the range of 10 to 100, depending on how large the peak energy stored in capacitor C is and how rapidly one wants to charge capacitor C. Lambda can be viewed as the number of energy slugs which the current pump must pump to fully charge capacitor C. Advantages of this novel design are its simplicity, its ability to operate synchronously with the discharge of the CD circuit, its inherent smooth rise in current and voltage which prevents false triggering of the discharge circuit 11 (especially under boost operation where energy must be delivered rapidly to the output circuit), and its insensitivity to the load, i.e. current pump 13 is insensitive to the state of the CD circuit 11 in terms of whether SCR 5 is OFF or ON since the transformed output capacitance (N**2)*C is much greater than Ce, and thus the input circuit is designed to work into what is a short circuit for practical purposes. In this generic form of current pump 13, element 26 may be a semiconductor switch or a diode, inductor 30 may be present or absent, and charging semiconductor element 29 may be present or absent (with capacitor 28 charging taking place through 31a when 29 is absent). Diode 33' shown across shunting FET switch 33 (shown with broken lines) is integral to the FET and is used to conduct the second half current cycle of the discharge cycle of capacitor 28. Output element 25 can be a range of components which will be governed largely by factors relating to how to most efficiently handle the power that is coupled to the output circuit during the capacitor 28 charging stage and the second half of the discharge stage. During these conditions voltage V2 is negative and without proper design would dissipate power or not contribute to charging of load capacitor 4. Ellipsis sections E in the circuit indicate room for optional placement (or absence) of various control filtering or power draw or limiting components. We note that the Circuit Controller is made up of blocks 16/19/20. We have tacitly assumed that the control of generic current pump of FIG. 1 corresponds to that given with reference to FIGS, la to 2e, and not to that of FIG. 4 where a gate clock oscillator 21 for triggering SCR 5 is needed. Controller 16, which includes an oscillator with waveform output shown in FIG. 2a, provides both the current pump control and the SCR 5 pulsing signal. Thus Controller (16/19/20) is very simple requiring only to perform the function referenced in FIG. 2a and to trigger the initial spark (which may be a ringing spark) upon receiving a trigger at input 18, and turning OFF the current pump 13 during the initial spark and restarting it at the end of the spark, and continuing as indicated in FIG. 2a. FIGS, la, lb, lc r and Id (only output stage shown) are preferred embodiments of the generic form of the current pump (FIG. I) in which inductor 30 is always present and switching element 26 is a diode 27a. FET switch 33, capacitor 28, and coil 31 perform the same energy transfer function already described with reference to FIG. 1. Charging time of capacitor 28 is deter¬ mined principally by values of inductor 30 (and capacitor 28). For a typical 50 uF value of capacitor 28, inductor 30 will have a value L (or L0) of about 60 uHenry for a charging period TC/2 of 180 usecs (leakage inductance of coil 31 is assumed to add about 10 uHenry in the cases of FIGS, la and lc). Such a charging rate corresponds to a power supply output of about 80 watts for 12 volts input (and noting that the charging is resonant or voltage doubling charging). Inductor 30 time constant Tl is very long, i.e. Tl = L/r = 50 uHenry/0.02 ohm = 2500 usecs for example, so that the short circuit current 10 (see FIG. 1) that flows while FET switch is ON for about 80 usecs (designated as 10) is limited by the inductance of inductor 30 (and not its resistance) to a value of 10 to 20 amps peak in this case. This corresponds to negligible resistive loss. An important feature of the operation of the current pump is that since FET switch 33 is turned OFF preferably near its peak second half cycle, then current 112 tends to cancel current 10 so the current through FET 33 is low or zero (negative) at TURN-OFF. Upon FET TURN-OFF, charging current 10 is now steered to charge capacitor 28, and instead of a zero initial charging current, a current has been established which will speed up charging of capacitor 28. FIG. la features the placement of a capacitor 25a and inductor 25aa (a reactive snubbing network) at the output of coil 31 to store and release (with minimum dissipation) energy during the charging stage of capacitor 28. Energy (of a negative voltage polarity) is coupled to the output circuit during the capacitor 28 charging stage because of the voltage drop across winding 31a, which equals the fraction Le/L of the total primary voltage VI, where Le is the primary winding leakage inductance given by: Le = Lpl * (1 - K**2) where Lpl is primary inductance of primary winding 31a, and K is the coupling coefficient of the coil 31; i.e. voltage drop across winding 31a generates current flow through the primary inductance 31a which couples power to the output winding 31b (in proportion o the primary leakage inductance Le of 31a to the inductance L of 30 as already stated). Snubbing network 25a/25aa will differ with each design, the objective being to experimentally pick those combinations of 25a and 25aa which maximize efficiency. In this figure is also shown a typical ignition output circuit including secondary winding 2 of a coil 3, and output capacitance 9 and spark gap 106. FIG. lb differs principally from FIG. la by the presence of a bypass charging element 29 (an SCR shown in this case). The main disadvantage here is the need for the extra component and the loss it represents during the charging stage. The advantage is that in bypassing inductor 31a we have eliminated coupling losses and eliminated the influence that coil 31 has on charging rate. In this case a diode 25b and resistor 25c (of say 50 ohms) are used at the output since current through diode 25b is. small. FIG. lc differs from FIG. la in its output stage, which incorporates a full wave charging output stage with center tapped secondary 31ba with two windings 31bb and 31bc. Winding 31bc may have a much larger number of turns than winding 31bb to compensate for the lower (input) voltage coupled to it from the primary winding 31a. This is done in order for winding 31bc to produce a sufficiently high output voltage to overcome the output potential 14a (developed by winding 31bb) so that it "sees" load capacitor 4 and is able to contribute some charge, albeit relatively small, to the load during the charging of capacitor 28, and even during the second half cycle of discharge of capacitor 28. Another useful construction of coil 31 is to have a tighter coupling between primary winding 31a and winding 31bc than between windings 31a and 31bb. This will provide the two benefits of a shorter second half discharge cycle of capacitor 28 and a corres- pondingly relatively higher output voltage (and higher second half cycle input current 112 for cancelling current 10 during the TURN-OFF of FET 33. This can be accomplished by winding the two secondary windings 3Ibb, 31bc on different arms or sections of a magnetic (preferably ferrite) core and distributing the primary turns between the two, with more of the primary windings wound colinear with winding 31bc for tighter coupling to it. However, it should be noted with reference to construction of output stage of windings of FIG. lc (and FIG. Id), which will be appreciated better with reference to FIGS. 2a through 2e, that one should preferably not fire the output CD circuit 11 (FIG. 1), if such a load circuit is connected, during the charging stage, (as is ideal in the cases of FIGS, la, lb) since in the present case one would be "shorting out" the supply during the time when winding 31bc delivers power. If such power delivery is negligible one may wish to ignore the "shorted output effect" during ignition firing and/or isolate the output from the short. However, prefe¬ rably, the current pump is operated at a higher frequency, say at 12 KHz instead of at 6 KHz, and is shut off, say, every fourth cycle for ignition firing of the CD circuit 11 (during an ignition firing made up of a sequence of pulses at 3 KHz) * Finally, in FIG. Id we show an output stage of the current pump with a full-wave bridge made up of diodes 32a, 32aa, 25b, 25bb, with circuit elements 25' and 25" defined with respect to the specific current pump design. For example, if the coil 31 turns ratio N is much higher than dictated by the voltage regula- tion setting for. normal operation, elements 25', 25" may be short circuits since the peak voltages across 31b for all stages of the current pump operation may be high enough to insure that 25b, 25bb also conduct when node 14b is of positive polarity. FIGS. 2a - 2e depict waveforms of a typical operation of the current pump during the ignition pulsing period, i.e. when the CD circuit 11 is fired say every 320 usecs as indicated in FIG. 2e (at a 3 KHz frequency). Rectangular wave oscillations of FIG. 2a (generated by circuit 16 of FIG. 1) turn FET 33 ON at the rising edge R, and turns FET 33 OFF after time T2 (and SCR 5 ON every other time) at its falling edge F; and turns FET 33 ON again Toff time later. FIG. 2b shows the expected current flow with the current slopes differing before and after F. FIG. 2c shows the expected FET 33 switching current 112 (FIGS. 1 - lc) which charges up capacitor 4. FIG. 2d shows the variation of voltage VI (or V12) with time; and FIG. 2e shows the discharge current 134 through capacitor 4 (which has an assumed value of 5 uFarads i.e. 4.5 - 5.4 uFarads as per convention. In all cases of operation, during the ignition firing period SCR 5 (FIG. 1) is triggered, and during the non-firing period it is not, but the current pump 13 continues to free-run during the non-firing period run until capacitor 4 is fully charged, which is controlled by information from voltage divider 45, 46 (FIG. 1). In FIG. 3a transistor switch 27 is connected in series with battery 10, capacitor 28, and primary winding 31a of coil 31. When transistor 27 is turned ON, capacitor 28, which is generally initially dicharged, is charged to near double battery voltage 2*VB. Output diode 25 allows current to flow in secondary winding 31b during the charging (while diode 32 is backbiased) to provide a short charge time Teh. given by: Teh. = pi * SQRT(Lpl*Ce) * SQRT(1 - K**2), where pi = 3.142, SQRT represents square root of, and "«*" indi- cates exponentiation. Lpl is inductance of winding 31a, Ce is capacitance of capacitor 28, and K is the coefficient of coupling of the coil 31. When the charging cycle is complete, switch 33 is activated and transistor 27 is simultaneously turned OFF. Switch 33 (which can be an NPN transistor, SCR, or FET as shown) discharges capa¬ citor 28 through winding 31a and transfers energy stored in capacitor 28 through diode 32 to output capacitor connected to terminal 14a. The energy transfer time (discharge time) is approximately equal to, and somewhat longer thari, Teh., since the coupling parameter K is now slightly lowered (from say 0.994 to A high efficiency coil 3 connected to 14a has a low primary inductance relative to a conventional coil and hence only a small effect on K and hence on discharge time relative to the charge time. A typical charging time for this circuit is 80 microseconds which can be attained with the following component values: Lpl = 1000 microHenry ("uH"); K = .994 Ce = 50 microFarad ("uF") N = 20 to 24 (N is the transformer turns ratio) Peak charging current Ich is around 25 amps, and peak discharging current 112 is in the same range but depends on the output voltage conditions according to: 112 = (VI - V20/N)/Z12 where VI is voltage on 28, and V20 is voltage on a load output capacitor connected to terminal 14a, both just prior to switching of switch 33, and Z12 is the source impedence of combination capacitor 28 and winding 31a given by: Z12 = SQRT(Lpl/Ce) * SQRT(1 - K**2) Power output is 100 watts and efficiency is 85% for a 2N5301 transistor and two parallel MTP25N05 FETs making up switches 27 and 33 respectively. This process of energy transfer continues until terminal 14a reaches a preset value of voltage sensed at junction 45a of output voltage divider 45/46. Point 46a is sensed to keep power supply OFF during the operation of discharge circuit 11. The voltage waveform VI follows a simple cosine function, with a corresponding simple sinusoid current waveform. Λ FIG. 3b represents the "synchronous" current pump (SCP) with "power boost" and is similar to the circuit of FIG.3a except for the addition of switch 29 (SCR shown), inductor 30, and the elimination of diode 25. In addition, switches 27 and 33 are larger to handle the higher peak currents of about 50 amps during boost operation. In its operation it is identical to that of FIG. 3a during the non-boost stage defined by the time when circuit 11 (FIG. 1) is inactive (point G of FIG. 4 is low) except that the charge time Tchs is longer to give a higher efficiency to compensate for the lower efficiency during the boost period. When level G is high (circuit 11 is active), switches 27 and 29 are ON (and switch 33 is OFF) for the periods T2 of FIG.2a, and switch 33 is ON (and switches 27 and 29 are OFF) during the OFF time Toff, shown in FIG. 2a. SCR 29 allows for rapid charging of capacitor 28 through inductor 30, which is chosen to give the desired boost charge time Tchb approximately equal to circuit 11 firing time T2. For example, for the following values: Ce = 100 uF Le = 8 uH where Le is inductance value of 30) Lpl = 160 uH N = 20; K = 0.99 gives: Tchb = pi * SQRT(Le * Ce) = 90 microseconds ("usecs") T12 = 50 usecs Tchs = pi * SQRT(Lpl * Ce) » 400 usecs where Tchb and Tchs are the times required to charge capacitor 28 during the boost time (G high) and the non-boost time (G low) respectively. T12 is the time required to discharge capacitor 28 and charge the output capacitor connected to terminal 14a. Tchs is longer than Teh because of the absence of diode 26 which presents an open output during the charging stage ' (for G low), i.e. there is no mutual coupling between the windings 31a and 31b, and hence the charging time constant is determined solely by inductor 31a in combination with capacitor 28. The peak boost currents Ichb, discharge current 112, and slow charge current Ichs are approxi¬ mately 40 amps, 50 amps, and 15 amps respectively based on the above values. Suitable components for switches 27, 29, and 33 for these peak currents are Motorola 2N5685 (50 amp NPN transistor), MCR265-2 (55 amp, 50 volt SCR), and either two MTH35N05 or three MTP25N05 in parallel. The energy stored and discharged during each charge transfer (sequential firing of switches 27 and 33) is: Esd = 1/2 * (Ce * (2VB)**2) = 30 millijoules for the above values assuming battery voltage VB of about 12 to 14 volts. These values translate to a non-boost power of 70 watts and a boost power of 200 watts (versus a constant output power of 100 watts for the example of FIG. 3a). With reference to the current pumps described in FIGS. 1 to 3b, we note that the peak voltage that capacitor 28 and FET 33 need to tolerate is twice the input voltage (although this may be higher if current 10 is not allowed to complete its charge (half) cycle before FET switch is activated and 10 builds up a DC compo¬ nent which can up to quadruple the input voltage on capacitor 28). The lowest practical rating of FETs is 50 volts, and of non- electrolytic (e.g. polypropelene) capacitors is 100 volts, i.e. one pays no less for these higher than needed ratings. Therefore, the current pump DC to DC converter is ideally suited for 24 volt applications, such as military vehicles, and commercial vehicles once the change is made to higher voltage batteries. In the above preferred embodiments it is assumed that the preferred operation of the current pump is one with a high degree of synchronization with the operation of a CD ignition i.e. one does not need an additional timing oscillator circuit 21 shown in FIG. 4. Therefore, in viewing FIG. 4, one must note that inclu¬ sion of oscillator 21 provides an additional degree of freedom which may not be necessary. Fig. 4 depicts a preferred embodiment of the invention (excluding the plug) incorporating the synchronous current pump (SCP) of FIG. 3b as the power supply means 13, and showing in detail a controller means 16 for controlling power supply means 13 in conjunction with discharge waveform generator composed of circuits 19, 20, and 21, described in U.S. Patent application SN 688,020 (which is incorporated herein by reference), wjth the ^ exception that in the present application, output of transistor 62 includes resistor 91 (instead of a diode 91) and additional NPN transistor 92 with resistor 93a at its output to hold gate 5a of "ringing" SCR 5 high for several discharge cycles of discharge circuit 11. In addition, the state of Gate Pulse Width controller 20 is used as an input to controller means 16 to signal it to operate in a normal (non-boost or idle) charging mode (G low), or in the high or "boost" mode (G high) for higher power output when the ignition is firing. Controller 16 for SCP 13 includes a low voltage reference including zener diode 36, resistors 36a, 36b, 36c and capacitor 36d which serves to establish the reference for voltage regulator 42 and establish switching thresholds for comparators 41 and 43; an astable multivibrator including IC 35, capacitors 35a and 35b, and resistors 35c and 35d, which determines the frequency of switching in the idle mode (gate G low); logic gates 47a, 47b, 47c (NOR gates) and comparators 41 and 42 which sequence the switching in the two modes of operation (G high and low); tran¬ sistors 37 and 39 and resistors 27b, 37a, and 37c which switch transistor 27 to conduct current from the battery to charge up capacitor 28 when transistor 39 is biased ON through resistor 39a; a shunt switch including FETs 33a and 33b and resistors 33c and 33d and C-mos drivers 34a and 34b, which shunts the current when 34a and 34b are activated by applying positive voltage to their inputs; a charge speed-up switch including SCR 29, resistor 29b, diode 29c, and capacitor 29d; a voltage divider including resistor 45 and 46; and a steering diode 48. In the idle mode (gate signal G Low), power is supplied by SCP through the following operation. Since voltage 14a is below the regulation point (for SCP to operate), output of comparator 42 is held low; output of C-mos logic gate 47b is forced to logic low (since G is low), which fixes output of comparator 43 and prevents it from triggering the charge speed-up switch 29. Ouput of logic gate 47c is the inverse of the output (pin 3) of astable multivibrator 35 and, as long as the rectified secondary voltage 14a stays below the set point, the output of logic gate 47a is the inverse of the output of logic gate 47c, and is thus the same as the output of 35. Thus the astable multivibrator causes the series switch 27 and shunt switches 33a/33b to alternate with one being ON while the other is OFF. The frequency of oscillation is chosen to achieve efficient charging and discharging of the LC " resonator 31a/28 and is determined by the natural frequency of the circuit (described with reference to FIG. 3b). When the voltage regulation set point is reached comparator 42 changes state causing its output to become high which forces the output of logic gate 47a to become logic low, which switches OFF series pass switch 27 and switches ON the shunt switch 33. Resistors 42a and 42b of comparator 42 constitute a network which creates a small amount of hysteresis in the voltage regulator loop and avoids the possibility of oscillation in it. Resistor 42c is the pull-up resistor for the open collector output of the comparator 42. In the boost mode (gate signal G at logic high), the output of logic gate 47c is forced low removing the astable multivibrator from the circuit. When rectified secondary voltage 14a is low i.e. less than a specified level of, say, 2.4 volts DC, steering diode 48 conducts causing the output of comparator 41 to become high and the output of logic gate 47b to become low. When output of 47b goes low, output of comparator 43 goes high, triggering gate 29a of SCR 29 and causing it to conduct which short circuits the primary of transformer 31 allowing for- rapid charging of capacitor 28. At the same time the output of logic gate 47a goes high which switches OFF shunt switch 33a/33b and switches ON series pass switch 27 which charges capacitor 28. When rectified secondary voltage 14a is high -(greater than or equal to 2.4 volts DC), series pass switch 27 is turned OFF and shunt switch 33a/33b is switched ON transforming energy to the secondary of transformer 31 and reverse biasing SCR 29 which turns OFF, and output of comparator 43 switches to a low state, which prepares trigger capacitor 29d for the next cycle. As in the idle mode, if rectified secondary voltage 14a reaches the set point series pass switch 27 is turned OFF, and shunt switch 33a/33b is turned ON, causing charge speed-up switch 29 to switch OFF. Comparator 41 with its resistors 41a, 41b, and 41c constitutes the circuit which instructs the controller as to the state of the output switches 5 and 6. When these switches are in the conducting state (G is high and hence output of logic gate 47c is low), the voltage at 14a is below 3 volts which is trans- mitted by diode 48 to the comparator 41. The high output from 41 maintains output of logic gate 47b low, which with 47c also low insures that switch 27 is conducting to recharge capacitor 28 while the capacitor 4 is delivering energy to the spark plug. Conversely, when the output switches are not conducting and the voltage 14a rises sufficiently to be sensed by comparator 41, the low output from 41 will turn OFF switch 27 and turn ON the shunt switch 33a/33b. The energy which has now been stored on capacitor 28 will be transferred through transformer 31 and deposited on capacitor 4 to partially recharge it. Resistors 41a and 41c insert a small amount of hysteresis in the circuit for stability, while resistor 41b is a pull-up resistor for the open collector of the comparator 41. Certain components in the circuit produce specific functions which are described below: SWITCH 27: Resistor 27b serves to assure rapid and reliable turn off of NPN transistor 27 when it is not biased ON. PNP transistor 37 is used to amplify the base drive to NPN transistor 27 with the lowest possible forward voltage drop (from the battery to the emitter of NPN transistor 27). Resistor 37a assures a rapid and reliable turn OFF of PNP transistor 37 when it is not biased ON. Resistor 37c biases PNP transistor 37 when NPN darlington tran- sistor 39 is saturated. Resistor 39a biases NPN darlington transistor 39 when the output of logic gate 47a is positive. As a result, series pass switch 27 is switched ON when the output of logic gate 47a is high. SHUNT SWITCH 33a/33b: Two FETs 33a and 33b are used as switching elements, and additional power FETs may be added to increase the efficiency of shunting of high currents. FETs 33a/33b are biased ON when a positive voltage is applied to their gates. C-mos inverter buffers 34a and 34b supply the drive to bias the gates of FETs 33a and 33b. Up to six inverter buffers may be connected in parallel to speed up the switching time of the FETs 33a/33b. When a positive voltage is applied to the inputs of buffers 34a and 34b, their outputs are driven to ground and FETs 33a/33b do not conduct. When a negative voltage is applied to the input of buffers 34a and 34b their outputs are driven to the positive supply voltage and FETs 33a/33b conduct. CHARGE SPEED-UP SWITCH 29: It includes SCR 29 which is normally biased off by resistor 29b, clipping diode 29d which prevents the reverse biasing of the gate anode junction of 29, and capacitor 29d which provides a positive voltage pulse to the gate 29a of SCR 29 each time the output of comparator 43 makes the transition from low voltage to high voltage. Resistor 43a is a pull-up resistor for the open collector. ZENER REFERENCE SUPPLY: The reference supply for the comparators 41, 42 and 43 comprises the zener diode 36 itself in conjunction with resistors 36a, 36b, 36c, and capacitor 36d. The zener diode voltage of 6.2 volts has been chosen for its inherent thermal stability but since this value is too high for use when the battery voltage drops to 6 volts, the zener stabilized reference voltage is divided by resistive divider 36b and 36c to a value of approximately 4 volts. Capacitor 36d smoothes the ripple of this reference voltage. Resistor 36a is the load resistor for the zener diode. -17- FIG. 5 depicts an optimized -simplified current pump, desig¬ nated hereinafter as OSCP, which has the same circuit configura¬ tion as the embodiment of FIG.la excepting for the removal of the inductor 25aa. Referring to FIG. 5 in which like numerals denote like parts (with respect to FIG.la), there has now been developed a precise prescription for picking the output capacitor 98 and the values of other parameters so as to optimize operation of the current pump and in effect provide a DC to DC power converter with what is believed is an unprecedented simplicity, with both a very high power output and efficiency, with a smooth operation, and a power output to input voltage characteristic which makes it ideal for automotive applications where the battery voltage may cover the range from fourteen volts down to seven volts under engine cranking conditions. A major breakthrough has been achieved through the OSCP by converting the (capacitor 28) charging stage of the current purap, which ordinarily places the wrong polarity at the output stage indicated by voltage V2, to one where the "wrong" or opposite polarity voltage is used instead to great advantage. Such "wrong" polarity voltage V2 is used to energize a capacitor 98 of capaci¬ tance C2, which in turn through precise definition is used to not only feed back a portion of its energy to storage capacitor 28, but to guarantee cancellation of the current through FET 33 during FET turn-off (eliminating voltage spikes), and to minimize the overall current pump working o ' scillat±on period. Before discussing the OSCP, an optimum turns ratio N of the transformer 31 can now be defined (especially with respect to automotive applications). It can be shown that the maximum energy transfer for the current pump occurs when the turns ratio N is equal to twice the output voltage V2 divided by the input voltage VI, i.e. when NO = 2 * V2/V1, where NO is the optimized turns ratio. For the present application, we expect VI to equal approximately 30 volts (for a fourteen volt battery), and assuming we desire to charge an output capacitor 4 (of say 4 to 8 uFarads value) to 360 volts, then the value of NO would equal: NO = 2 * 360/30 = 24. Such a design is ideal since it allows capacitor 28 to be charged to full voltage rapidly when the engine is running (and. VI is 14 volts), and slowly when the engine is being cranked and there is much time available and the voltage is low (VI is 8 volts). For a recently built OSCP whose parameters will be disclosed by means of an example, the time required to charge an 8 uFarad capacitor to 350 volts was 6 msec at approximately 14 volts, and 60 msec at approximately 8 volts (representing the time that is typically available at cranking conditions). Further before discussing the OSCP, a factor is presented which has bearing to the discussion of the OSCP. This relates to the equation which governs the voltage to which capacitor 28 is charged from the battery connected to point 9. It was already disclosed that this voltage is twice battery voltage VB, but that is only the case if the initial current is zero. On the other hand an initial current 100 for charging capacitor 28 is set up as a result of current flow 10 through inductor 30 when FET switch 33 is turned on and then off (where time T2 represents the time of this current build-up). This initial current 100 produces two important results with respect to charging of energy capacitor 28. It reduces the charge-up time Teh and increases the voltage to greater than twice the voltage (difference). For example, if 100 is equal to the maximum current Im that would ordinarily flow in the absence of an initial current 10, then the capacitor 28 charge time Teh is reduced by 25% (from Ϊ80 to 135 degrees of the total charge time oscillation period), and the peak voltage VI appearing across capacitor 28 is raised frow 2 to 2.4 times battery voltage VB (assuming that capacitor 28 is initially uncharged). In fact, voltage VI for an initial current 100 is given by: VI = VB * [1 + SQRT[1 + (I00/Im)**2]] For example, for a voltage of 12.5 volts available for charging capacitor 28, i.e. the voltage to the right of diode 27a which is preferably a schottky diode such as an MBR745, VI is 30 volts if 100 is equal to Im. Hence, the OSCP disclosed herein is designed to advantageously use these benefits, and particularly to increase the power output of the OSCP. -19- The operation of the OSCP is best understood with reference to FIGS. 5a through 5d, where FIGS. 5a, 5b, 5c correspond to FIGS. 2a, 2b, 2c respectively, which have already been disclosed and described. The periods of oscillation indicated in the figures all have the same time scale and thus correspond to each other as indicated. The key aspect of the OSCP is the placement of capa¬ citor 98 at the output, with a value C2 given approximately by: C2 = (N**2) * CO With this value of C2, it can be shown that: T22 » .7 * T12 where T12, shown in FIG. 5b, has already been defined as the time for capacitor 28 to discharge through its first half cycle and charge capacitor 4 through diode 32. T12', equal to T22, is the second half cycle of discharging of capacitor 28 which now charges capacitor 98 to the opposite polarity. T22 is the half-period of oscillation of capacitor 98 producing current 122' as indicated in FIG. 5 and discharging through transformer 31 and the primary circuit defined by capacitor 28, FET 33 (which is switched ON), and transformer winding 31a, to produce current 122 as shown in FIG. 5 and FIG. 5d. Starting with the charging phase, capacitor 28 is charged to a voltage VI and capacitor 98 is charged in phase to an opposite polarity voltage V22 within the time period Toff as indicated in FIG. 5a, which corresponds to the FET 33 o f-time Toff. When FET 33 is switched ON, indicated by the rising edge R of the rectan- gular waveform of FIG. 5a, capacitor ,98 discharges producing the oscillating current 122 shown in FIG. 5d, which for the particular values indicated earlier flows through the FET integral diode 33a with a value 1220 near its maximum value at or just beyond the zero current crossing point of 112. The current 1220 adds to the reverse current of 112, indicated as 1120'in FIG. 5c, to more than cancel the initial charging current 100 (FIG. 5b), where 100 is given by: 100 = VB * T2/L0 An immediate result is that an independent means has been provided to force the current through the FET 33 to zero at its turn-off. A second very important result is that the current 122 is in the direction to further charge capacitor 28 during its charging stage indicated by Toff. With reference to FIG. 5a through 5d, the current pump cycle period Tcp (equal to T2 plus Toff) is shown to equal 105 usecs. The period T12 is shown as 50 usecs (and T22 approximately as 35 usecs according to our prescription). The determination for the FET 33 OFF period can be made by inspection of FIG. 5d, where the objective is to fit an integral number of oscillation periods T22 within the total period Tcp. Inspection of the waveforms indicate three T22 periods which include a very short duration spike 1223 at the end of the second half cycle. This occurs because as capa¬ citor 98 begins its third half cycle discharge FET 33 is off and diodes 33a and 27a are back biased, so the path the current must take in the primary circuit is through the very small capacitor 99 of capacitance value C3 of the order of magnitude 0.01 uFarad. This period is very short relative to T22 so that capacitor 98 flips its charge in effect instantaneously and proceeds with what is in effect its third period, which has the same direction as the second period and delivers charge to the energy capacitor 28 through diode 33a. The FET 33 swith-ON time is then set to correspond to approximately three half cycles of T22, or three times 35 usecs or 105 usecs as indicated.. The peak initial current at FET 33 turn-OFF 100 is controlled by choke inductor 30 and duration T2 and is preferably set to correspond to the maximum charge current Im. Clearly, 100 can be set at another value, say a higher value to produce a higher initial voltage VI of, say, 40 volts or three times the available battery voltage; this condition would reduce the required charge time Toff from 180 to 120 degrees. However, whatever value of L0 is used it must also be chosen to correspond to the remaining time Toff available for charging capacitor 28 . The total (180 degree, zero initial current) charge time Teh is given in this case by: Teh - pi * SQRTCL0 * CO * MU/(1 + MU + L0/Lpl)] where pi = 3.142 MU = (N**2)*C2/C0 -21- For the preferred value of MU = 1, Teh reduces to: Teh = pi * SQRT[L0*C0/(2 + L0/Lpl)] which for L0 much less than Lpl further reduces to: Teh = pi * SQRT[L0*C0/2] Hence, it is seen that once a capacitor CO has been selected and a transformer 31 designed (with among other things an optimized turns ratio NO) then capacitor 98 is determined; then period T12 is measured, period 3*T22 calculated (which defines period Tcp), and periods T2 and Toff taken as approximately 60% and 40% of Tcp respectively. Inductor 30 is then selected to provide a charge time Teh less then that defined above by an amount related to the initial current 100. It should be noted that since capacitor 98 charges in phase (or 180 degrees out of phase depending on one's view) with capacitor 28, then preferably the charge time*Tchf as shown in FIG. 5b should be equal to or just greater than Tcp as already defined. This is because at the end of Tchf capacitor 98 has in effect been reset (recharged) and will begin to disharge. But according to the present prescription FET 33 should have just been turned-ON so that capacitor 98 sees capacitor 28 as ' its load through transformer 31 so that the quoted relationships between T22 and T12 are adhered to. Finally, it has been determined that from at least a size and cost perspective it is preferable to use a smaller capacitor CO than earlier disclosed, say in the range of 5 to 20 uFarads. Clearly this will result in higher frequency operation, typically in the range of 10 to 20 KHz, but this is not a problem given the state-of-the-art of ferrite cores and other parts. In fact, the time periods T2 (60 usecs) and Toff (45 usecs) shown in FIG. 5a correspond to a 20 uFarad energy capacitor 28. More specifically, these times correspond to the following design: CO = 20 uFarads; NO = 25; Np = 24; Ns = 600; C2 = 0.03 uFarad; C3 * 0.01 uFarad; C = 8 uFarads;. L0 = 40 uHenries; Tchf = 60 usecs; Tchf = 3/4 * Tchf = 45 usecs = Toff V2 = 360 volts; VI = 30 volts; Np, Ns are the primary and secondary turns of the transformer 31 respectively, and Tchf has been defined as the time required to charge the capacitor 28 with an initial current 100 corresponding to Im, i.e. it is 135 of the normal 180 degrees, or 3/4 of Tchf. It is ofcourse tacitly assumed here that L0 and the other para¬ meters are selected such that all these conditions are satisfied. In the end, one experimentally makes the fine adjustments for a given capacitance CO and transformer to obtain .the properly phased operation of the OSCP. While it may not be apparent here it has been experimentally shown that the OSCP represents a real breakthrough in DC to DC poer converters, especially for automotive applications. It is extremely simple, being made up of seven components, an input and output diode, an input and output capacitor (neglecting capacitor 99), one choke inductor, one transformer, and only one active component, namely FET 33, which for a 60 to 100 watt automotive power supply is preferably made up of two parallel low voltage (50 volt) Motorola FETs, such as the recently developed, very high efficiency (low RDS) BUZll FETs. It is low cost; it is very efficient; it can be easily scaled for a wide range of power levels; it provides rapid charging at high voltage and slow charging at low (engine cranking) voltage; it is insensitive to the load (can tolerate short circuits); and has other advantages which will become apparent to those familiar with power supplies, especially for automotive applications. Since certain changes may be made in the above apparatus and method without departing from the scope of the invention herein involved, it is intended that all matter contained in the above description, or shown in the accompanying drawings shall be inter- preted in an illustrative and not in a limiting sense.

Claims: CLAIMS What is claimed is: 1. A Simplified Current Pump or SCP DC to DC power converter which includes in series with a low voltage power supply inductor means, energy storage and switching capacitor means, and the primary winding of a transformer, and further including means defining a shunt switch to ground with first diode means across it, wherein said shunt switch is connected to the point between said inductor means and said capacitor means, . and second diode means connected between the output of said transformer and an electrical load " . 2. The system defined in claim 1 wherein an inductor and a high voltage capacitor are connected- in series across the output of said transformer between the high side of said transformer and anode of said second diode and the ground side of output of transformer. 3. The system defined in claim 2 constructed and arranged so that low voltage input power is converted to a higher voltage output power by sequentially turning said shunt switch ON and OFF with a duty cycle no greater than 50%, where duty cycle is defined as the ratio of the switch ON time t-ON to the sum of the OFF time t-OFF and ON time t-ON, defined also as total time t-TOT, expressed as a percentage. 4. The system defined in claim 3 wherein said switching capacitor has a capacitance CO and the time period t-ON corresponds to the sum of T-SH1 plus 1/2 of T-SH2, where T-SH1 is the first half discharge cycle of the switching capacitor (CO) discharging through said transformer via shunt switch , and T-SH2 is the second half of the disharge cycle or recharge half-cycle which recharges CO through said transformer via said first diode. 5. The system of claim 4 wherein said inductor has an inductance value LO and resistance R0 and t-ON is much less than the time Tl or inductive time constant LO/RO and t-TOT is approximately equal to the charge time Teh which is the time required to charge CO through LO with zero initial current, which is approximately equal to half the period of oscillation defined by CO and LO. 6. The system defined in claim 5 wherein said shunt switch comprises at least one FET switch (SS) and wherein the load connected to output of said transformer is a capacitor "Cl". 7. The system defined in claim 6 wherein Cl is a capacitor of a capacitor discharge circuit including a transformer and high voltage switch (SHV) for producing and delivering high voltage to a high voltage output circuit (HVOC). 8. The system defined in claim 7 wherein HVOC is an ignition output circuit including at least one spark plug defining at least one spark gap for producing ignition sparks and switch SHV is an SCR. 9. The system defined in claim 8 wherein said SCR is fired sequentialy to produce multiple spark pulses in phase with swit- ching frequency of FET switch SS wherein said SCR is turned ON and OFF during OFF time t-OFF of said FET switch SS. 10. The system defined in claim 9 wherein the SCR turn-ON moment corresponds essentially to the FET switch SS turn-OFF moment. 11. The system defined in claim 10 wherein SCR ON time corres- ponds to the FET switch OFF time. 12. The system defined in claim 11 wherein the energy storage factor Eo is in the range of 5 to 50 millijoules, where Eo is defined as Eo = 2 * CO * (Vo**2). 13. The system defined in claim 12 wherein Vo is a twelve volt battery and CO is between 20 and 100 microfarads. 14. The system defined in claim 13 wherein L0 is in the range of 20 to 200 microhenries, Cl is in the range of 2 to 10 uFarads, and transformer turns ratio N is between 16 and 32. 15. The system defined in claim 14 wherein the charge time Teh is in the range of 40 to 400 useconds. 16. The system defined in claim 12 wherein Vo is a twenty four volt battery and CO is between 10 and 100 ufarads. 17. The system defined in claim 12 wherein Vo is a forty eight volt battery and Co is between 5 and 50 ufarads. 18. The system defined in claim 1 wherein an additional shunt switch is connected between intersection of capacitor CO and primary of transformer and ground. 19. The system defined in claim 18 wherein said additional shunt switch is an SCR and wherein a diode and resistor is connected across the output of the transformer. 20. The system defined in claim 1 wherein the secondary winding of said transformer is a center tapped secondary winding with each output containing a series diode, said outputs being connec- ted together to a load capacitor. 21. The system defined in claim 20 wherein the two secondary windings defined by the said center tapped secondary winding are of unequal number of turns. 22. The system defined in claim 21 wherein the major of the two secondary windings, defined as the winding performing the major or main power transfer versus the minor winding, is the winding with the lesser turns of said two secondary windings. 23. The system defined in claim 22 wherein the ratio of the number of turns of the major to minor winding is approximately equal to the ratio of the peak voltage across the primary winding during the charging of the switching capacitor and the value of the voltage of the fully charged switching capacitor. 24. The system defined in claim 20 wherein the minor secondary winding of said center tapped secondary is more closely coupled to the primary winding than is the major secondary winding coupled to the primary winding. 25. The system defined in claim 24 wherein the second " discharge time period T-SH2 is between one quarter and one half the first discharge time period T-SH1 of the two periods making Λ up the energy transfer discharge cycle of the switching capacitor CO switching and discharging through the primary transformer ' winding. 26. The system defined in claim 24 wherein said minor secondary winding has more turns than the major secondary winding. 27. The system defind in claim 26 constructed and arranged such that said two secondary windings of said transformer are wound on different parts of the core of said transformer and more turns of the primary winding are wound collinear with said minor secondary winding to produce a tighter coupling to said minor winding. 28. The system defined in claim 1 wherein magnetic sense of the primary and secondary windings are opposed, i.e. when voltage on primary winding is positive, voltage on secondary is negative. 29. The system defined in claim 1 wherein output of said trans- former has connected to it a full wave bridge consisting of four diodes. 30. The system of claim 29 wherein other circuit elements are connected at output of secondary of said transformer in conjunc- tion with said four diodes of said full wave bridge. 31. Electric circuit for producing a both high ignition voltage and high ignition current discharge, said current discharge in- eluding VHF-UHF capacitive current and lower frequency inductive current, said circuit comprising: (a) a low turns ratio high efficiency ignition coil with low primary and secondary resistances and a turns ratio less than eighty, (b) at least one discharge capacitor connected to said spark coil and having a with capacitance greater than 3 microfarads; (c) at least one switch for discharging said capacitor into the primary of said spark coil; and (d) a high efficiency DC-DC converter power supply of the current pump type connected for powering said capacitor. 3S. Electric circuit of claim 31 wherein said power supply includes in series with a low voltage power supply, a solid state εferies switch, a low voltage energy storage capacitor, and the primary of a transformer, and further including a shunt switch to ground connected to the point between said solid state series switch and said low voltage capacitor, and a first diode connect- ng the output of said transformer to said discharge capacitor for charging said discharge capacitor. 33. Electric circuit of claim 32 wherein said low voltage energy storage capacitor has a capacitance between 50 and 250 ufarads. 34. Electric circuit of claim 33 wherein said low voltage power supply is a 12 volt car battery and said discharge capacitor is 400 volt capacitor, and wherein said transformer has a turns ratio between 16 and 32. 35. Electric circuit of claim 34 wherein turns ratio of said transformer is between 20 and 25. 36. Electric circuit of claim 34 wherein said solid state series switch is a power transistor and said shunt switch is one or more FETs in parallel. 37. Electric circuit of cLaim 34 including a second diode con- nected across the output of said transformer with anode to ground. 38. Electric circuit of claim 34 wherein said power supply is constructed and arranged to be a synchronous current pump by the inclusion of a small inductor between said battery and the intersection point of said shunt and series switches, and inclu— sion of a boost solid state switch across the input of said transformer. 39. Electric circuit of claim 38 wherein said boost switch is an SCR with cathode connected to ground. A 40. Electric circuit of claim 39 constructed and arranged so that the time constant to charge said low voltage energy storage capacitor through said small inductor and said boost switch ' corresponds to the inductive discharge oscillation period of the discharge circuit connected at output of said transformer, and time required to discharge energy storage capacitor through said transformer and charge output discharge capacitor corresponds to the time between spark pulses of the discharge circuit formed primarily by said discharge capacitor and ignition coil. 41. Electric circuit of claim 31 including members defining a spark gap of a width of at least 0.080 inches between them, said members being connected across the secondary of said spark coil for producing a spark current discharge. 42. Electric circuit of claim 41 wherein said members defining a spark gap constitute a said plug being mounted in a combustion chamber of an internal combustion engine. 43. Electric circuit of claim 42 wherein said IC engine contains at least one movable member and wherein said spark plasma dis- charge is made between said plug and movable member. -29- 44. Electric circuit of claim 43 wherein said spark plug has a tip which is essentially converging to a point about 1/8 inch diameter and wherein said movable member also has converging tip such that there is produced a focussing of the electric field in the gap wherein said spark plasma discharge is produced. 45. The system defined in claim 1 wherein the magenetic sense of the primary and secondary windings of said transformer are of an opposed sense and wherein a high voltage capacitor of capacitance C2 is connected across the secondary winding of said transformer between the high side of said secondary winding and anode of said second diode and the ground side of the winding, wherein the value of C2 is approximately equal to C0/N**2, where CO is capa- citance of said energy storage and switching capacitor and N is said transformer turns ratio, and wherein the current pump swit- ching time Tcp corresponds to approximately three times capacitor C2 half oscillation period T22, and period Tcp is divided approximately in a ratio of three to two representing said shunt switch ON time and OFF time respectively.

Inventors: WARD MICHAEL A V, WARD, MICHAEL, A., V.
Applicants: CUMBUSTION ELECTROMAGNETICS, COMBUSTION ELECTROMAGNETICS, INC.
Application year: 1987
Main classification: H02M3
IPC codes: F02P3, H02M3, H03K3