Power on demand beam deflection system for dual mode crt displays

Abstract: A cathode ray tube beam deflection system operable in slew and random stroke and periodic raster display modes provides automatic power supply voltage switching (l6, l8) to maintain linear operation and high efficiency. Control of automatic switching is obtained by continuously monitoring yoke voltage, yoke current, and deflection voltage, a power supply voltage being switched to a voltage of higher magnitude to provide a higher deflection rate when the yoke voltage exceeds a predetermined level at a predetermined current polarity and returned to a power supply voltage of lower magnitude when the higher deflection rate is no longer required.

Description: The invention relates generally to electromagnetically deflected beam display systems and more particularly to power supply control circuits for providing linear operation and high efficiency in random stroke and periodic raster display modes and during slew of a cathode ray tube electron beam. The power efficiency of deflection systems that display both raster and stroke writing is relatively low due to the inductive deflection yoke and the high driving voltages required for magnetic deflection to assure adequate writing speed. Sophisticated airborne navigation displays with increased display area and information content require a significant increase in power consumption, while space and available power is limited. Since the deflection yoke driving circuit consumes a significant portion of the total display power, the power efficiency of the deflection system may be greatly enhanced if the required driving voltages can be reduced. Since the rate of deflection for a raster display is generally much higher than for stroke deflection, the supply voltages applied for raster deflection are correspondingly higher. To obtain maximum slew speed during the stroke display also requires a relatively high supply voltage or reduced yoke inductance, both of which increase power dissipation of the system. However, during the writing phase of the stroke display, relatively low voltages may be satisfactory. Hence it is desirable to switch the power applied to the system to provide the minimum voltage required to assure linear operation. One form of prior art apparatus providing dynamic power reduction was disclosed in US-A-3,965,390. This involved monitoring the yoke voltages so that the power supply voltage is switched to a higher voltage when the yoke voltage exceeds a predetermined level and returned to the power supply voltage of lower magnitude when the higher voltage is no longer required. This invention utilised a flyback raster for retrace and provided reduced supply voltage only during the stroked deflection period when reduced writing speed was allowable. An improved system as described in co-pending European Patent Application No. 87302922.7 (Publication No 0,245,941), which constitute prior art according to Article 54(3) EPC, provides an external raster/stroke control signal from a symbol generator to selectively apply a plurality of power supply sources to a push-pull yoke driver amplifier in accordance with the displayed mode of operation. Efficiency was further enhanced during raster operation by applying a control signal derived from the voltage developed across the yoke to synchronise the power switch closures. However, the limited voltage available during the stroke period resulted in inadequate high speed slewing capability. Further, it was desirable to eliminate the need for an external raster/stroke control signal in order to minimise the complexity of the display circuitry. The present invention describes a system for optimising power conservation during the raster and stroke displays while permitting increased slewing speed. The invention is controlled by internal signals developed in the yoke driver amplifier without the need for external control signals. Since the internal switch control signals do not discriminate between stroke and raster operation, stroke writing efficiency is optimised even at high slewing speeds. Moreover, minimum power dissipation is also obtained during slewing conditions by varying the applied yoke driver voltages to that required to obtain linear operation. The present invention is defined in the appended claims and provides a deflection amplifier for a cathode ray tube employing a magnetic deflection coil to position the beam of a cathode ray tube along its face. The amplifier comprises a differential amplifier, a feedback element, a deflection amplifier, a plurality of voltage sources, a preamplifier, and a plurality of switches. The differential amplifier responds to beam positional signals and to a feedback signal representative of the current through the deflection coil. The error signal thereby developed is coupled to drive the preamplifier, which in turn causes the deflection amplifer to provide a current proportional to the input signal to the deflection coil. The switches are connected to the voltage sources to selectively and independently supply the deflection amplifier with sufficient current to maintain linear operation in raster, stroke, and slew modes of operation while minimising power consumption. Control signals for activating the switches are derived by sensing the voltage developed across the deflection coil and the current flowing therethrough. By applying one of the voltage sources to the deflection amplifier when a first voltage level is developed across the deflection coil, and switching to a second voltage source when a second voltage level is developed across the deflection coil, independent of the display mode and dependent only on the rate of change of current in coil; power consumption is minimised while providing high rates of deflection speed. The present invention will now be more particularly described, by way of example, with reference to the following drawings, in which:- Figure 1 is a functional block diagram of the apparatus of the present invention, Figures 2A and 2B are simplified schematic circuit diagrams of a preferred embodiment of the present invention. Figure 3 is a diagram with input and output waveforms for a sinusoidal deflection signal applied to the present invention. Figure 4 shows input and output waveforms for a triangular deflection signal useful in understanding the operation of the present invention. Figure 5 is a diagram illustrating input and output waveforms at a high writing speed. Referring to Figure 1, a power on demand electron beam magnetic deflection system operable to provide linear deflection in the stroke mode for random deflection of the beam and while slewing the beam, and in the raster mode for periodic deflection of the beam, includes a differential amplifier 1O, a preamplifier 12, a push-pull amplifier stage 14, a deflection yoke 20 mounted on a cathode ray tube (CRT) (not shown), and a yoke current sampling resistor 22. A positive power switch 16 coupled to receive current from a plurality of power supplies +15V, +45V, and -15V receives control signals from preamplifier 12 on line 24 and energises push-pull amplifier 14 on line 28. A negative power switch 18 receives current from -15V, -45V, and +15V power supplies and control signals from preamplifier 12 via line 26, and provides current to push-pull amplifier 14 on line 30. An input signal V IN , representative of the desired beam deflection, which may be in stroke mode, raster mode, or during slewing of the beam, is applied on line 36 to the non-inverting input of differential amplifier 1O. A feedback signal V FB , derived by sensing a voltage drop across resistor 22 proportional to yoke current I O , is provided on line 38 to the inverting input of differential amplifier 1O. The two signals are algebraically subtracted and amplified in differential amplifier 1O to provide an error signal V e on line 40 which is coupled to the input of preamplifier 12. Preamplifier 12 provides an amplified voltage V I for driving push-pull amplifier 14. Amplifier 14 operates in a conventional manner to provide an output signal V O on line 42 for driving a magnetising current I O through deflection yoke 20. The current I O also flows through series connected line 32 to sampling resistor 22 to develop a feedback signal V FB . The signal V FB is proportional in magnitude and polarity to the current I O . When impressed on differential amplifier 1O in a closed loop manner, with linear amplifier operation, the resultant current I O is directly proportional to V IN . In operation, a deflection signal V IN is applied to differential amplifier 1O to develop an output signal V e . Signal V e is amplified by preamplifier 12 to provide a driving signal V I to push-pull amplifier 14. Amplifier 14 provides an output signal V O to energise deflection yoke 2O. The current I O flowing in yoke 2O is sampled in series resistor 22 to develop a feedback signal V FB which is proportional to the current I O . Differential amplifier 1O algebraically combines V IN and V FB to develop resultant signal V e . This signal drives the preamplifier 12 and push-pull amplifier 14 in closed loop fashion so that the current waveform I O replicates the deflection signal V IN . In accordance with the driving voltage V O required to generate a desired yoke current, which in turn is a function of the writing speed, power switches 16 and 18 are individually energised to select one of a plurality of power supplies in accordance with substantially the minimum supply voltage required to assure linear operation. For the positive power switch 16, which energises yoke 20 when positive deflection current is demanded, a control signal on line 24 from preamplifer 12 energises power switch 16. This control signal is responsive to the deflection command V IN on line 36 and to the feedback signal V FB on line 38. The magnitude of signal V O is sensed and communicated to switch 16 through amplifier 14. The combination of these signals determines which of the supplies coupled to switch 16 will be made available on line 28 to push-pull amplifier 14. The operation of negative power switch 18, which energises yoke 20 when negative deflection current is commanded, follows in a similar manner to energise the lower section of push-pull amplifier 14 in response to control signals on lines 26 and 30. Figure 2 illustrates a schematic circuit diagram of a preferred embodiment of the invention. Not shown are conventional circuit elements used to enhance the frequency response, increase transistor current gain, and stabilise the system. Input stage 1O comprises a conventional differential amplifier coupled to receive the beam deflection signal V IN on line 36 at one input and a feedback signal V FB developed across resistor 22 and coupled at node 56 to a second input on line 38 to sample the current passing through deflection yoke 20. The output of amplifier 1O is an error voltage V e which is applied on line 40 to current amplifier 11 of preamplifier 12. Current amplifier 11 draws current from a +15V supply through transistor Q1 and from the +45V supply through transistors Q2, Q7 and Q8. Amplifier 11 draws current I₁ at pins 1 and 2 from emitter 15a of transistor Q1. Amplifier 11 is further energised at pins 7 and 6 from a-15V supply through transistor Q9 and from a -45V supply through transistors Q1O, Q11, and Q12. The output 4 of amplifier 11 is coupled to load resistor 13, which is connected to ground at reference numeral 9. Coupled between the collectors of transistors Q2 and Q1O are series connected diodes CR3-CR8 which provide predetermined bias voltages V B , V C , V D and V E . Current amplifier 11 is a unity gain buffer, such as type LHOOO2 as manufactured by National Semiconductor Corp., Santa Clara, CA. The cathode of diode CR3 is coupled to the anode of diode CR4. The cathode of diode CR4 connects at node 47 to base 57b of transistor Q5 and to the anode of diode CR5. The cathode of diode CR5 is coupled to the anode of diode CR6 and the cathode thereof connected at node 49 to the anode of diode CR7 and the base 59b of transistor Q6. Diode CR7 has its cathode connected to the anode of diode CR8. A positive voltage source of +15V at terminal 56 is applied to the base 15c of transistor Q1. Transistor Q₁ draws current I₃ from transistors Q₂ and Q₈. Transistors Q₂, Q₇, and Q₈ are connected in a PNP Wilson Constant Current Source configuration such as is commonly employed in operational amplifier microcircuits. The base 17c of transistor Q2 is coupled to the collector 21b of transistor Q8 and the collector 15b of transistor Q1 at node 23. Emitter 17a of transistor Q2 and collector 19b of transistor Q7 are coupled at node 25 to the base 19c of transistor Q7 and base 21c of transistor Q8. Emitters 19a and 21a of transistors Q7 and Q8, respectively, are connected in common at node 27 to a positive high voltage supply at terminal 70, typically +45V. Collector 17b of transistor Q2 is coupled to the anode of diode CR3 and the cathode of diode CR2 at node 24. Pins 6 and 7 of amplifier 11 are coupled to supply current I₂ to emitter 31a of NPN transistor Q9. The base 31b of transistor Q9 is coupled to a -15V power source. Transistors Q1O, Q11, and Q12 are connected in an NPN Wilson Current Source configuration. The collector 31c of transistor Q9 is coupled to base 33b of transistor Q1O and collector 37c of transistor Q11 at node 35. Emitter 33a of transistor Q1O is coupled to collector 41c and base 41b of transistor Q12 and also coupled to base 37b of transistor Q11 at node 39. Emitters 37a and 41a of transistors Q11 and Q12 are coupled at node 43 to a -45V power supply. The collector 33c of transistor Q1O is coupled at node 26 to the base 61b of transistor Q4, the cathode of diode CR8, and the anode of diode CR9 of the negative power switch 18. The positive power switch 16 is comprised of transistors Q3 and Q13 and diodes CR1, CR2, CR11, CR13, and CR14, and coupled to +15V, -15V, and +45V power supplies. The +45V power supply at terminal 70 is coupled at node 27 to the anode of a constant current unidirectional conducting element CR1 such as type IN5314, as manufactured by Motorola Semiconductor Corp. The cathode of diode CR1 connects at node 45 to the base 53b of transistor Q13 and the anode of diode CR2. The cathode of diode CR2 is coupled at node 24 to the anode of diode CR3, the collector 17b of transistor Q2 and to the base 55b of transistor Q3. The collector 53c of transistor Q13 is connected to a +15V voltage source at terminal 68. A diode CR13 has its anode coupled to the emitter 53a of transistor Q13 and its base coupled to node 65. A diode CR14 has its anode coupled to a -15V power source at termianl 66 and the cathode coupled to nodes 65 and 67. Emitter 55a of transistor Q3 is coupled to the anode of diode CR11. Node 67 is coupled to the cathode of diode CR11 and to the collector 57c of transistor Q5. A +45V supply at terminal 711is coupled to collector 55c of transistor Q3. In a manner similar to positive power switch 16, negative power switch 18 is comprised of transistors Q4 and Q14, diodes CR9, CR1O, CR12, CR15, and CR16, and coupled to power sources supplying +15V, -15V, and -45V. The cathode of diode CR9 connects at node 57 to the base 63b of transistor Q14 and the anode of a constant current unidirectional conducting element CR1O. The cathode of element CR1O connects at node 43 to the -45V power source at terminal 76. Emitter 63a of transistor Q14 is connected to the cathode of diode CR15 and collector 63c to a -15V power source at terminal 74. Collector 59a of transistor Q6 connects to the anodes of diodes CR12, CR15 and CR16 at node 54. The cathode of diode CR12 is coupled to emitter 61c of transistor Q4. Collector 61A of transistor Q4 is connected to a -45V power source at terminal 69. The cathode of diode CR16 is connected to a +15V power source at terminal 72. Node 51 is connected to base 63b of transistor Q14. Push-pull amplifier 14 is comprised of diodes CR5 and CR6 and cascaded transistors Q5 and Q6 whose common emitter junction at node 52 is connected via lead 42 to energise deflection coil 20. Node 47 of the diode chain connects via lead 46 to the base 57b of transistor Q5. Emitter 57a of transistor Q5 connects via node 52 to emitter 59c of transistor Q6 and to one end of deflection yoke 20. Node 49 of the diode chain connects to base 59b of transistor Q6. The second end of deflection coil 20 is connected at node 56 to sampling resistor 22 and by line 38 to input the negative of differential amplifier 1O. Sampling resistor 22 is terminated to ground at reference numeral 58. In operation a signal V IN applied to differential amplifier 1O will result in a current I O proportional thereto in yoke 20. Thus, a positive-going signal applied to lead 36 will result in a positive yoke current, and a negative-going signal applied to lead 36 will result in a negative current in yoke 20. Assuming zero initial conditions, with a positive voltage V IN applied to differential amplifier 1O, a positive error voltage V will be applied to current amplifier 11. Power is drawn in the direction shown by arrow I₁ from the emitter of transistor Q₁ to pins 1 and 2 of current amplifier 11. Transistor Q₁ acts to buffer current amplifier 11 from the high voltage power sources. Collector current I₃ of transistor Q₁ is substantially equal in value to emitter current I₁. Transistors Q₇ and Q₈ are a matched pair configured as a Wilson current source and provide a current output I₅ at transistor Q₂ which is equal in magnitude to the current I₃ but oppositely polarised. Amplifier 11 also supplies idle current at pins 6 and 7 to buffer transistor Q₉. Thus, the output current I₄ at the collector of Q₉ is equal to the input current I₂ from pins 6 and 7 of amplifier 11 flowing to emitter 31a of transistor Q₉. A current I₆ at the collector 33c of transistor Q₁₀ is drawn through the diode chain CR₂-CR₉ and is equal in magnitude to the idle current I₄. Thereby when error voltage V e is equal to zero V₁ is approximately equal to O V and current I₅ = I₆. As signal V e becomes more positive, the current I₅ will increase relative to the current I₆. I₅ increases proportional to V e while the idle current is substantially constant. Accordingly, the voltage V₁ will increase positively. Conversely, as signal V e goes negative, the current I₆ will become greater than current I₅ and the output voltage V₁ becomes negative. In addition to providing output voltage V₁, preamplifier 12 provides bias voltages V B , V C , V D and V E , determined by the predetermined diode voltage drops across CR3-CR8. In operation, with power supplies of ±45 V, the output voltage V₁ will range over approximately ±41.5 V. The function of power control switches 16 and 18 is to supply the collectors of the output transistors Q₅ and Q₆ with the lowest supply voltage that will permit maintaining linear operation. Thus, one of the +45 V, +15 V, or -15 V supplies is selected by the positive power control circuitry and one of the -45 V, -15 V, or +15 V supplies is selected to supply negative output current to the collector of transistor Q₆. The sequential operation of the power control switches may be readily understood by consideration of an example. Since the amplifier 14 is driving an inductive load 20, the following polarity conditions for amplifier output voltage V O and yoke current I O will exist: Note that unlike a resistive load, a negative output voltage must be developed for positive output current and vice versa under some conditions of operation. All positive output current I O is supplied by the positive power switch 16, and all negative output current is provided by the negative power switch 18. The power control circuitry will select the lowest supply voltage as a function of the required electron beam deflection rate. The actual magnitude of the power supplies which are selected by the power switches is a function of the deflection rate of the input signal V IN . For illustrative purposes, a sine wave input signal may be selected for V IN , which will exercise a deflection amplifier of the type shown in Figure 2 over a writing rate up to approximately 236 in/sec on a 6″ x 6″ CRT face with 48° on-axis deflection angle. Figure 3 shows the output voltage waveform V O required to obtain an output current I O that is a replica of V IN . A sine wave input with a period of 8O µS is chosen for ease of analysis and to illustrate exercising both positive and negative control circuitry. It is assumed that a peak voltage of 1 V is applied. With sine wave input, the rate of change of current through the yoke ranges from O A/sec to 230 KA/sec. The output voltage V O corresponding to the applied deflection voltage to obtain an output current I O that is a replica of the applied deflection voltage V IN can be calculated as follows: V IN = sin (7.85 x 10⁴t) (1) where L = inductance of yoke (180 µh) dI O /dt = rate of change of output current with respect to time I O = yoke current (amp) R Y = yoke resistance (0.6 ohm) R S = sample resistor (0.34 ohm) Since the output current I O is forced to be proportional to the deflection voltage V IN by the feedback loop, the magnitude may be found from the relationship: |I O | = V IN /R S (3) Since R S has a typical value of 0.34 ohm, I O has a peak value of ±2.94 amp, hence I O = 2.94 sin (7.85 x 1O⁴t) (4) Neglecting the voltage drops across R Y and R S and substituting equation (4) and the value for L in (2) yields: V O = 41.5 cos (7.85 x 1O⁴t) (5) By considering the losses due to diode and transistor voltage drops and the resulting bias relationships, a table may be constructed which provides the minimum supply voltage required to generate the desired V O waveform. This is shown in Table 2 below. The effect of the yoke and sampling resistor voltage drops may be readily observed by considering a linear waveform, as in Figure 4. A sawtooth voltage V IN of one volt peak value is applied as the deflection waveform. The output waveform V O to obtain a yoke current I O that is a replica of V IN is found from equation (2). Assuming a deflection writing rate of 35 Kin/sec and a deflection sensitivity of 3.1A for centre to edge deflection on a 6″ x 6″ display, V O = (180 µh) (35 Kin/sec) (3.1A/3 in) + I O (0.6 + 0.34 ohms) (6) For positive deflection as shown at point 130 V O = 6.51 + 0.94 I O (7) For negative deflection as at point 132 V O = -6.51 + 0.94 I O (8) The current I O is I O = (±35 kin/sec) (3.1A/3 in) = (±36.17 KA/sec)t (9) Substituting (9) in (7) and (8) yields V O = 6.51 + 34 x 10³t (10) and V O = -6.51 - 34 x 10³t (11) For a deflection period of 171 µs, this results in a peak deflection amplitude of ±12.32 V. Referring to waveform V O at point 134, it may be seen that the effect of increasing yoke current is to increase the voltage drop due to series resistances R Y and R S , hence requiring an increasing yoke voltage V O . Referring to Table 2, it is seen that when I O is positive and V O between 9.41 V and 12.32 V, the +15V supply will be applied; when I O is positive and V O between -6.51 V and -9.41 V, the +15V supply is applied. When I O is negative and V O is between -9.41 V and -12.32 V, or I O negative and V O is between +6.5 V and +9.41 V, the -15V supply is applied. Thus, at the reduced writing speed, the system automatically selects the lowest voltage supplies. In operation, the required supply voltage will be a function of the desired output voltage and the polarity of output current, which in turn depends on the yoke inductances and rate of deflection of the electron beam. Figure 3 shows a family of waveforms corresponding to a sinusoidal deflection voltage V IN . Curve V IN shows a sine wave with amplitude 2 V peak-to-peak. The time base is divided into six intervals 1OO, 1O2, 1O4, 1O6, 1O8 and 11O, each interval corresponding to the utilisation of a particular power supply. While six supplies have been chosen for illustrative purposes, this is by way of example only and in principle the number of supplies may be extended or diminished. Corresponding to the deflection voltage curve V IN is the curve V O of the output voltage across deflection coil 2O. Since the coil is primarily inductive, the output voltage is shifted in phase by 90° in relation to the current I O . As an example, for the desired deflection on the CRT, a peak-to-peak amplitude of 93 V is required. The current waveform I O is in phase with the deflection voltage V IN by virtue of the feedback circuitry which forces the current waveform to be identical to the deflection voltage. The yoke current is scaled for a peak-to-peak value of 5.88A, which corresponds to a peak current of 2.94 A. Table 2 identifies the power supply voltage applied for each of the six intervals. Referring now to Figure 3 with continued reference to Figure 2, the operation of the positive power switch 16 will be considered in detail. Positive power switch 16 selects substantially the lowest supply voltage required to provide the desired output voltage V O . During interval 1OO the output voltage V O ranges between +41.5 and +13.4 V. Transistor Q₃ and diode CR11 are biased into conduction while transistor Q₁₃ and diode CR13 are not conducting. Diode CR14 is back biased and not conducting. Diode CR2 is back biased and not conducting. Thus, transistor Q₃ and diode CR11 conduct the output current from the +45 V supply at terminal 71 while the current paths from the +15 V and -15 V supplies are interrupted. Diode CR1 essentially provides a constant current source and isolation of loading effects on the +45V supply. Consider now interval 1O2 of Figure 3. The output voltage V O is seen to range between +13.4 V and -17.1 V. Over this range, the voltage at node 65 will vary between -15.7 V and +14.1 V. Diodes CR11 and CR14 will be biased for nonconduction over substantially the entire range. The voltage at node 45 varies from -14.3V to +15.5V, while at node 65 it varies between -15.7 V and +14.1 V, so that transistor Q₁₃ is biased for conduction. Diode CR13 is forwarded biased so that the output current I O is supplied by the +15 V supply at terminal 68. The voltage at collector 57c of transistor Q₅ will be between 0.7 to 1.4 V above the output voltage V O and therefore transistor Q₅ is always kept out of saturation. Considering now the operation of transistor Q₃ and diode CR11, during interval 1O2 the voltage applied between nodes 24 and 67 is insufficient to bias the components to conductivity. Therefore, transistor Q₃ and diode CR11 will be nonconducting for output voltage V O ranging from -17.1 to +13.4 V. When the output voltage V O is between -41.5 V and -17.1 V as in interval 104, diode CR14 will be biased for conduction. Assuming a typical diode voltage drop of 0.7 V, the voltage V F at node 65 will be -15.7 V. Similarly, considering the diode voltage drops for CR3, CR4 and transistor Q₅ the voltage V B at node 24 appearing at base 55b of transistor Q₃ will be V O + 2.1 V. Therefore, for V O between -41.5 V and -17.1V, voltage V B at node 24 will range between -39.4 V and -15V. It may be seen then that the voltage difference between nodes 65 and 24 will range between -23.7 V to 0.7 V. Since this voltage must be at least 1.4 V to forward bias diode CR11 and transistor Q₃ transistor Q₃ is turned off for V O ranging between -41.5 V to -17.1 V. Similarly, by counting diode drops for diodes CR2, CR3, CR4 and transistor Q₅ it may be shown that the voltage difference between nodes 45 and 65 will range from -23 V to +1.4 V. Therefore transistor Q₁₃ will be turned on when the voltage difference applied between the base 53b of transistor Q₁₃ and the cathode of diode CR₁₃ equals 1.4 V, and thus will be turned off for values of V O less than -17.1 V. For V O ranging from -41.5 V to -17.1 V diode CR14 conducts output current I O from the -15 V supply at terminal 66 while diodes CR11 and CR13 are back biased and therefore not conducting current. Hence, the +45V and +15V supplies are disconnected. The operation of the system during intervals 106-110 is similar except that negative power switch 18 will be operative in a similar manner to that of the positive power switch described above. Thus, during interval 110 the yoke voltage V O ranges between +41.5 to +17.1 V and is energised by the +15 V supply at terminal 72 acting through diode CR16 and transistor Q₆. Diodes CR12 and CR15 will be biased to nonconduction so that the -15 V supply at terminal 74 and the -45 V supply at terminal 69 do not provide output current. During interval 1O8 wherein V O ranges between +17.1 and -13.4 V, power is supplied by the -15 V supply at terminal 74 through diode CR15 and transistor Q₁₄ and Q₆. Diodes CR9, CR12, and CR16 are reverse biased. Finally, during interval 106 where V O ranges between -13.4 and -41.5 V, transistor Q₁₄ and diodes CR15 and CR16 are in a nonconducting state, while diode CR12 is forward biased, so that current is supplied from the -45 V supply at terminal 69 through diode CR12 and transistor Q₄ to transistor Q₆. It may be seen that the greater the rate of change of deflection voltage the higher the value of the power supply required. This can be shown by an additional example using a writing speed of 180 Kin/sec. Referring now to Figure 5, V IN represents a triangular waveform with a peak value of 1 V. The corresponding deflection yoke current I O is also a triangular waveform of peak amplitude 2.94 A whose magnitude has been determined as described above. It may be seen that the voltage waveform V O describes a ramp increasing from 33.48 V to 39.3 V and decreasing from -33.48 V to -39.3 V. The intervals 112, 114, 116, 118 of Figure 5 designate time intervals corresponding to operation of the power switching circuitry. Choosing the V baseline for the beginning of the control sequence designated by line 120, V IN is O V, I O is O A, and V O is 36.4 V. The positive voltage V IN applied to amplifier 1O results in a positive voltage V₁ at the cathode of diode CR5. Bias V H = 15.5V applied to the base 53b of transistor Q₁₃ and 37.1 V applied to the cathode of CR₁₃ through diode CR11 and transistor Q₃, results in reverse biasing transistor Q₁₃ and diode CR13 by a value of -21.6 V. Since the voltage at the anode of diode CR14 is -15V, and V F applied at node 65 to the anode of diode CR14 is 37.1 V, diode CR14 is reverse biased. Therefore no current flows from the -15 V power supply at terminal 66. Since positive current is being supplied and can only flow through the upper transistor Q₅ of push-pull amplifier 14, transistors Q₆, Q₄, and Q₁₄ are nonconducting. Transistor Q3 is turned on by the positive bias V B resulting from the positive signal V IN applied to amplifier 1O. Thus, output current is provided from the +45 V supply at terminal 71 through transistor Q₃ and diode CR11 to transistor Q₅ and deflection coil 20. This is consistent with Table 2 for positive yoke current. The diodes and transistors remain in the same state throughout interval 112 while the output voltage V O and the output current I O continue to rise as shown in Figure 5. At the end of interval 112, denoted by line 122, the output voltage V O has reached a value of 39.3 V and yoke current I O is at a peak value of 2.94 A. In order to provide the decreasing yoke current shown by region 114, the output voltage must be immediately reduced to -33.48V. Amplifier 1O senses the change in deflection voltage V IN and causes V₁ to decrease until V O has reached a value of -33.48V. Since I O is still positive, although decreasing, transistors Q₆, Q₄, and Q₁₄ remain in a nonconducting state. However, the state of the positive power switching circuitry changes as follows: transistor Q₃ and diode CR11 are turned off because of the high negative bias appearing at node 24 coupled from the output voltage V O , allowing for the diode voltage drops in CR3, CR4 and Q₅; the voltage V B at the base 55b of transistor Q₃ is approximately -31.4 V. Since diode CR14 clamps V F to -15.7 V, and since voltage V B and base 55b of transistor Q3 is -31.4 V, diode CR11 and transistor Q3 are back biased. Transistor Q₁₃ and diode CR13 are back biased because the voltage at node 45 and base 53b of transistor Q₁₃ is -30.7 V, while the voltage at node 65 is -15.7 V. Since diode CR14 is biased for conduction, the output current I O is supplied from the -15 V power supply terminal 66 and controlled by transistor Q₅. These conductive states continue through interval 114. At the end of interval 114, denoted by line 124, the output voltage V O is continuing to decrease while V IN reaches a value of O V and I O has a value of OA. At this point, entering interval 116, the output current I O changes in polarity from positive to negative. Therefore, transistor Q₅ and diode CR14 no longer conduct current and the output current is provided through transistors Q₄ and Q₆ and diode CR12 from the -45 V supply at terminal 69. Diode CR16 is reverse biased by the negative voltage V G applied at anode junction 54, which has a value of approximately -37.1 V, and the +15 V supply at the cathode. A negative potential of -15.5 V appears at node 51 and is applied to the base 63b of transistor Q₁₄, while V G = -37.1 V is applied to the anode of CR15, so that NPN transistor Q₁₄ and diode CR15 are non-conducting. Therefore no current flows from the -15 V supply at terminal 74. These conductive states continue throughout interval 116. At the end of interval 116, denoted by line 126, the yoke voltage V O has reached a value of -39.3 V, the output curent I O is at a value of -2.94 A, and the deflection voltage V IN is -1 V. Since V IN now commences to increase in a positive direction, V O must rapidly change from -39.3 V to a value of +33.48 V in order to provide the required increase in yoke current. Since the output current I O is negative at this point, transistors Q₃, Q₁₃, and Q₅ remain nonconducting. However, as V O increases, negative power switch 18 changes state in the following manner. The positive voltage of 33.48 V developed across yoke 20 results in biasing diode CR16 to be conductive and supplies current I O from the +15 V supply at terminal 72 through transistor Q₆. Transistor Q₄ and diode CR12 are reverse biased by the positive voltage V E - V G applied to node 26 with respect to node 54, so that the -45V supply is disconnected. Transistor Q₁₄ and diode CR15 remain nonconducting because of the positive bias V I -V G applied between nodes 51 and 54. Therefore no current is provided by the -11 V supply at terminal 74. The foregoing conditions continue through interval 118. At the end of interval 118, the output current I O increases to positive polarity. Therefore transistor Q₆ and diode CR16 stop conducting current while transistors Q₃ and Q₅ and diode CR11 are biased for positive conduction. Diode CR14 is reversed biased by the positive voltage V F = 37.1V applied from transistor Q₅ to node 65 and the negative -15V supply at the anode. Therefore no current flows from the -15V power supply at terminal 66. Transistor Q₁₃ and diode CR13 remain nonconducting because of the negative bias V H - V F = -21.6 V applied between nodes 45 and 65. This completes a full cycle of operation. It should be noted that for this mode of operation (180 Kin/sec) transistors Q₁₃ and Q₁₄ remain off for the entire cycle and current does not flow through diodes CR13 and CR15. It may be seen from Table 2 that since the output voltage V O is not required to develop values in the range of -17.1 V to +13.4 V for positive I O and +17.1 V to -13.4 V for negative I O the plus and minus 15 V power supplies are not required and transistors Q₁₃ and Q₁₄ are not exercised. Conversely, if the writing speed is decreased to ±35 Kin/sec, as in the earlier example, transistors Q₁₃ and Q₁₄ and the ±15 V power supplies are adequate to supply the current throughout the cycle and therefore transistors Q₃ and Q₄ and diodes CR11, CR12, CR14 and CR16 remain nonconducting. When the writing speed is increased to, for example, 180 Kin/sec, then the ±45 V power supplies will be required. It may be seen from the foregoing that the invention provides the following advantages: a. High power efficiency by applying substantially the minimum power supply voltage necessary to assure linear operation. b. Automatic switches to provide the minimum power level consistent with the deflection rate. c. Minimises power dissipation in both raster and stroke modes of operation. d. Provides high rate slew rate capability during stroke writing. e. Does not require auxiliary control signals and associated circuitry.

Claims: An electron beam magnetic deflection amplifier for a display system controllably operable to provide deflection in a stroke mode for random deflection of the beam, a raster mode for periodic deflection of the beam, and a slew mode for traversing the beam at a maximum deflection rate, the amplifier comprising: input means (10) responsive to an input signal (V IN ) indicative of a desired deflection of the beam, for providing an output signal (V E ) responsive to the input signal (V IN ); preamplifier means (12) comprising a buffer amplifier (11) responsive to the output signal (V E ), current source means (Q₈, Q₇, Q₁₁, Q₁₂) for providing an output current indicative of the magnitude and sense of the output signal (V E ) and, responsive to the output current, for providing a further output current opposite in sense to the output current, a plurality of cascaded diodes (CR3-CR8) providing predetermined voltage drops and coupled to receive the output current and the further output current, for providing a plurality of predetermined bias voltages and a variable bias signal responsive to the input signal for commanding a power amplifier means (14); a power amplifier means (14) having first and second cascaded sections (Q₅,Q₆), coupled to receive particular ones of the bias voltages, for applying current to a deflection coil (20) operatively coupled to the electron beam, and for providing a desired deflection in accordance with the sense and rate of change of the input signal (V IN ); a plurality of voltage sources (66,68,70,72,74,76); a plurality of transistor switch means (16,18) coupled between said plurality of voltage sources and said first and second cascaded sections, biased by associated diode means (CR1,CR2,CR11,CR13,CR14; CR9,CR10,CR12,CR15,CR16) with predetermined voltage drops for operation in a non-saturated switching mode and for selectively applying voltage sources of positive and negative polarity to said first and second cascaded sections, responsive to the output current and the further output current from the preamplifier means (12) and also responsive to a voltage derived from the voltage (V₀) developed across the deflection coil (20) and a voltage drop across one of the first or second sections of the power amplifier means (14), the derived voltage following the voltage (V o ) across the deflection coil (20), a predetermined one of the switch means (16,18) being activated for a predetermined polarity of the deflection current when the derived voltage attains a first predetermined magnitude and polarity and deactivated when the derived voltage attains a second predetermined magnitude and polarity, the first section of the power amplifier means (14) being coupled to one of the switch means (16,18) for energising the electron beam in a first predetermined direction and the second section being coupled to another of the plurality of transistor switch means (16,18) for energising the electron beam in a second predetermined direction; and said plurality of voltage sources (66,68,70,72,74,76) being of further predetermined magnitudes and first and second polarities, a pair of said voltage sources of equal magnitude and opposing polarity being coupled to each one of said plurality of transistor switch means, and a further pair of said voltage sources of a given magnitude with opposing polarities being coupled, respectively, to ones of the plurality of transistor switch means (16,18), whereby a voltage source of sufficient magnitude is provided to the power amplifier means (14) which allows sufficient current to flow through the deflection coil (20) to accomplish the desired rate of change of beam deflection while maintaining linear operation of the power amplifier means (14) and minimising power consumption thereof, independent of the mode of operation of the display system. An amplifier according to claim 1, characterised in that the power amplifier means comprises a push-pull amplifier (14) with the first and second sections comprising cascaded transistors (Q₅,Q₆), each of the transistors having a base electrode (57b,59b) for receiving a control bias from the preamplifier means (12) and an emitter electrode (57c,59c) coupled in common and to the deflection coil (20), one of the transistors (Q₅,Q₆) having a collector coupled to one of the plurality of switches (16,18) and a further one of the transistors (Q₅,Q₆) having a collector coupled to a further one of the plurality of switches (16,18). An amplifier according to claim 2, characterised in that the switch means (16,18) comprises a first transistor (Q₁₃,Q₁₄) having a base (53b,63b), a collector (53c,63c), and an emitter (53a,63a) electrode, the base being coupled to a source of constant current (CR1,CR10) and to one of the cascaded diodes (CR2,CR9), the collector being coupled to one of the plurality of voltage sources (68,74) of a further predetermined magnitude and first and second polarities, the emitter (53a,63a) being coupled to first diode means (CR13,CR15), the first diode means being energised in response to sums of the derived voltage and the biases applied to the base electrode, the first diode means being coupled to second and third diode means (CR11,CR14;CR12,CR16), the first, second, and third diode means (CR11,CR13,CR14;CR12,CR15,CR16) being coupled for unidirectional current conductivity to one of the collectors (57a,59a) of the first and second cascaded transistors (Q₅,Q₆), the second diode means (CR14,CR16) being coupled to receive a further one of the voltage sources of further predetermined magnitude and first and second polarities; and a second transistor (Q₃,Q₄), having a base (55b,61b), an emitter (55a;61c), and a collector (55c;61a) electrode, the base (55b;61b) thereof being coupled to a further one of the cascaded diodes (CR3,CR8), whereby a predetermined voltage differential is maintained between the first mentioned and the second mentioned base electrodes (53b,55b,63b,61b), the collector (55c,61a) of the second transistor (Q₃,Q₄) being coupled to receive a still further one of the voltage sources of further predetermined magnitude and first and second polarities, and the emitter (55a,61c) of the second transistor (Q₃,Q₄) being coupled to energise the third diode means (CR11,CR12) in response to the bias voltages and the voltage drops. An amplifier according to any of claims 1, 2 or 3, characterised in that the preamplifier means (12) further comprises first output means (4) coupled to a load resistance (13), terminal means carrying control currents (I₁,I₂) in the current source means, (Q₈,Q₇,Q₁₁,Q₁₂), the terminal means being coupled to an emitter electrode (15a,31a) of a transistor (Q₁,Q₉) also having base (15c,31b) and collector (15b,31c) electrodes, the base being coupled to a power source, the collector being coupled to the current source means; the current source means comprising a pair of transistors (Q₈, Q₇, Q₁₁,Q₁₂) having base (21c,19c,37b,41b), collector (21b,19b,37c,41c) and emitter (21a,19a,37a,41a) electrodes, the emitter electrodes of said pair being coupled in common to a further power source, the base electrodes (21c,19c;37b,41b) of said pair being coupled in common to the emitter (17a,33a) of a further transistor (Q₂,Q₁₀) having base (17c;33b) collector (17b,33c) and emitter (17a,33a) electrodes, the collector electrode (15b,31c) of the first mentioned transistor (Q₁,Q₉) being coupled to the base electrode (17c,33b) of the further transistor (Q₂,Q₁₀) and to a first collector electrode (21b,37c) of the transistor pair (Q₈,Q₇,Q₁₁ ,Q₁₂) emitter (17a,33a) of the further transistor (Q₂,Q₁₀) also being coupled to a second collector electrode (19b,41c) of the transistor pair (Q₈,Q₇,Q₁₁,Q₁₂), the collector of the further transistor (Q₂,Q₁₀) being coupled to the cascaded diodes (CR3-CR9), whereby the terminal means provides a first predetermined current proportional to the output signal to the first mentioned transistor (Q₁,Q₉), and the collector (17b,33c) of the further transistor (Q₂, Q₁₀) provides a second predetermined current in a sense opposing said first predetermined current to the cascaded diodes (CR3-CR8). An amplifier according to any of claims 1 to 4, characterised in that the input means further comprises a differential amplifier (10) having first and second inputs, the first input being responsive to the input signal (V IN ); and further comprising an impedance (22) connected in series with the deflection coil (20) for providing a voltage (V FB ) representative of a current (I₀) flowing therethrough and fed back to the second input for comparison with the input signal (V IN ), for deriving an error signal (V E ) indicative of the difference between the input and feed back signals (V IN ,V FB ) for controlling the current (I₀) supplied by the power amplifier means in linear operation. An amplifier according to claim 5, characterised in that the first input comprises a non-inverting input (36) and the second input comprises an inverting input (38).

Inventors: WEINDORF PAUL F L, WEINDORF, PAUL F.L
Applicants: HONEYWELL INC, HONEYWELL INC.
Application year: 1987
Main classification: H03K4
IPC codes: G09G1