Apparatus for radio-frequency generation in resonator tank circuits

Abstract: A method of and apparatus for exciting harmonic generation of radio frequency in a tank resonant circuit (L, L c , C) by bipolar spaced, half-cycle pulses, wherein the period of the tank resonant frequency oscillations is 1/N the period between successive pulses of similar polarity, where N is an odd integer greater than unity.

Description: The present invention relates to apparatus for the generation of radio-frequency pulses or continuous waves in antenna resonator circuits or similar tuned circuits through excitation by sequential pulses, being particularly useful for the generation of bursts of radio-frequency energy as for Loran-C or similar radio navigation or other pulses, or for continuous waves in resonant loads or similar systems. The generation of such high power radio frequency pulses by successive positive and negative half-cycle d.c. pulses for such navigation pulse generation is described, for example, in U.S. Letters Patent No. 3,889,263; and such generation may be effected by a plurality of SCR-controlled sequential inverters connected with magnetic pulse compression circuits feeding the common high-Q antenna load as in U.S. Letters Patent Nos. 3,711,725 and 3,786,334. While such systems work admirably well in practice, it has now been discovered that similar operation can be successfully obtained with much fewer required sequential inversions and otherwise less costly apparatus, and/or with excitation by fewer d. c. pulses or half-cycles that are relatively narrow and widely spaced, instead of broad and more contiguous as in the prior systems. Additionally, this technique is well suited for the generation of high power CW transmissions, particularly in the VLF band as, for example, 50 to 30 kHz. Underlying the invention is the discovery that a generator of output pulses of fixed width can be caused to deliver useful harmonic energy to a resonant circuit and its load over a range of resonant frequencies, simply by insuring that the pulse-repetition frequency and the resonant frequency are appropriately harmonically related. This enables the use of a single generator or a set of generators in a rotating or sequential time sequence to obtain more power and a smoother output wave in the resonator with the rather surprising result of a lack of a constraining relationship between the pulse half-cycle width and the resonator half-cycle width. An object of the present invention, therefore, is to provide a new and improved apparatus for such sequential pulse excitation of radio-frequency resonator, tuned or tank circuits that reduce the excitation by widely spaced successive half-cycle pulses more suited, indeed, in the large current pulse, relatively infrequent switching capabilities of SCR switching circuits. A further object is to provide a novel radio-frequency generator and technique wherein harmonic energy is delivered to a resonator circuit and its load by bipolar pulses or the like of fixed width, and over a range of resonant frequencies, by ensuring appropriate harmonic relationship of pulse repetition frequency and the resonant frequencies and without the constraint of a relationship between the pulse half-cycle width and the resonator circuit half-cycle width. An additional object is to provide an improved sequential pulse excited tank circuit radio-frequency generator of more broad application, as well. Other and further objects will be explained hereinafter and more particularly pointed out in the appended claim. According to the invention there is provided apparatus for generating radio frequency pulses or continuous waves comprising a power generator for generating sequential half-cycle pulses of alternately opposite polarity and applying the same to a resonant load, said power generator having first and second saturable inductor switching controls each comprising an SCR and a saturable inductor connected in series, the saturable inductors being connected together to form a common node, a filter capacitor interconnecting the SCRs, two series-connected pulse capacitors, being connected in parallel with the filter capacitor, an output transformer having a primary winding connected between the junction of the said pulse capacitors and the said common node of the saturable inductors, the secondary winding of the output transformer being connected to the resonant load, said load comprising a tank resonator circuit provided with tuning means for tuning the frequency of said tank resonator circuit so that the tank resonator circuit oscillates at a predetermined frequency upon excitation by said power generator, said predetermined frequency being a multiple of the repetition frequency of said half-cycle pulses, a centre-tapped DC supply source, first and second inductors connected between the positive terminal of the DC supply source and the second SCR and between the negative terminal of the DC supply source and the first SCR, respectively, first and second series-connected switching diodes connected across the DC supply source, with the junction between the switching diodes connected to the centre tap of an autotransformer arranged in series between the junction of the said pulse capacitors and the said centre tap of the DC supply source so that when the voltage appearing at the junction between the pulse capacitors is above a predetermined positive threshold, the second switching diode conducts to divert the current from the autotransformer to the DC supply source, first and second step-up windings, each being magnetically coupled to a respective one of said first and second inductors, said step-up windings being connected in series to each other and to a third diode, said series-connected arrangement being connected across the DC supply source, said step-up windings providing a voltage step-up ratio with respect to the total voltage appearing across the first and second inductors, said third diode being conductive when the voltage across the filter capacitor exceeds a predetermined threshold, thereby diverting the current from the first and second inductors to the corresponding step-up windings and thus terminating the charging of the filter capacitor, so as to prevent simultaneous conduction of the SCRs from shorting out the DC supply source, and a voltage sensing circuit connected across said filter capacitor for inhibiting said SCRs in response to the existence of said excess voltage across the filter capacitor. The invention will now be described with reference to the accompanying drawings wherein: Fig 1 is a circuit diagram of a preferred embodiment of the invention; Fig 2 is an oscillogram of the bipolar pulses produced by preferred SCR-controlled discharges of the excitation pulse generator of Fig 1; Figs 3A and 3B are respectively oscillograms of steady state and resonator build-up waveforms generated in the circuit of Fig 1; Fig 4 is a graph showing optimum operation found to be achievable with appropriate pulse period (and width) and tank circuit resonant period adjustments in accordance with the invention; and Fig. 5 is an oscillogram showing the resonator current and effective clamping against the condition of the excitation energy exceeding the rate of harmonic energy utilization in the tank resonator circuit. In Fig. 1, the resonant load or tank circuit is shown in the form of a tapped inductance L-L c resonated with a capacitor bank C at a particular critically selected frequency, later described, with the total series loss resistance R s determining how much antenna current I a can be circulated with a given source power. Adjusting the total mutual inductance L c and M provides a means for matching a given tank (or resonator) loss resistance to the capabilities of the generator at a particular frequency. The excitation of this tank circuit is from a VLF sequential inverter circuit G of the SCR pulse generating type, also later described, which is coupled to energize the tank circuit with successive current pulses in a burst. At the beginning of a burst, the VLF source delivers its pulses into a resonator without circulating energy, and the load resistance presented to the VLF inverter is virtually zero. The resonator circulating current and the coupled load voltage faced by the VLF source build up exponentially toward their steady-state levels with a time constant τ =Q/π f o (where Q is that of the resonator or tank circuit including the output load; f o , the resonant frequency excited therein), as does the apparent load resistance. An important feature of this generator is its current-source behavior during the interval required to fill up the resonator or tank circuit. The sharply tuned resonator will extract energy near f o from any harmonic-rich current wave passed through the coupling inductor L c . In the source described here, L c is made a part of the discharge inductance in a push-pull SCR pulse generator, and receives a stream of bipolar (say, 35 µ s-wide) half-sine or current pulses (i.e. alternately opposite polarity pulses). The ability of SCRs to switch narrow, high-current pulses efficiently at kHz rates is thereby exploited, without encountering the associated device limitations on recovery time and dv/dt. Between drive-current pulses, the resonator supplies VLF power to the loss resistance R s , analagous to Class C vacuum-tube circuit operation. A clean sine wave will result if the resonator decay time τ is much longer than the interval between pulses. In tests over 10 to 30 kHz, successful operation was obtained with Q=200 and repetition rates from 3,000 to 6,000 pulse-pairs per second. Good d.c.-to-VLF conversion efficiency requires careful control of pulse-generator losses, since all the drive-current harmonics contribute dissipation, while only the harmonic at f o delivers useful output. Typically, 80 kw of resonator power for 100 kw of d.c. input was obtained in these tests. Referring to the specific circuit details of the SCR pulse generator high-power circuits of the resonator exciter G, Fig. 1, the power generator is symmetrical about a horizontal center line, having upper and lower symmetrical series-connected SCR₁-L₁ and SCR₂-L₂ saturable inductor switching controls. To understand how the pulse generator G operates, suppose that the center-tapped d.c. source (½E dc on a side) has been connected directly across the filter capacitor C₃, shunted by equal capacitors C₁ and C₂, and where C₃>>C₁+C₂, and ignore the clamping and latch-up protection circuits, so-labelled to the left, for the present. Assume that an initial positive charge has been placed at the junction of pulse capacitors C₁ and C₂, so that the voltage e c at their junction is positive with respect to ground E. SCR₁ is then forward-biased, while SCR₂ is back-biased. When SCR₁ is gated on, a half-cycle sinusoidal resonant discharge of C₁ occurs through the primary of an output transformer T₁ and the saturable magnetic inductor L₁, where the transformer primary connects from the junction of capacitors C₁ and C₂ to the junction saturable inductors L₁ and L₂. Simultaneously, the series combination of capacitor C₂ and the d.c. power supply (shunted by C₃) discharges along the same path, which adds energy from the supply to the circuit. After half a discharge cycle, the current through SCR₁ reverses and SCR₁ turns off (pulse I in Figs. 2 and 3B) leaving a negative voltage e c at the junction of C₁ and C₂ Now SCR₂ is forward-biased, SCR₁ is back-biased, and a small resonant-frequency current portion (Ia₁ in Fig. 3B) circulates in the tank. Once sufficient time has elapsed for SCR₁ to recover to its insulating state, SCR₂ can be gated to produce an oppositely directed discharge pulse (pulse II, Figs. 2 and 3B) through the primary of T₁ via inductor L₂. This time it is C₂ that has the direct discharge path, and C₁ whose current pulse passes out of the d.c. supply. Coherent reinforcement of the resonator oscillation will result if the second pulse occurs an odd number (greater than 1) of tank-frequency half-cycles after the first; i.e., if the frequency f o is an odd multiple of the pulse repetition frequency. Succeeding discharge pulses will encounter ever larger levels of opposing voltage at f o on the primary of T₁, due to passage of the circulating current through L c +M. Steady state is reached when the energy lost at f o in the load R s between pulses is just balanced by the energy pulse put into the generator from the d.c. supply during each discharge pulse. Fig. 2 shows the forward voltage across SCR₁ and the pulses of discharge current I and II through the primary of T₁, taken during steady-state operation on f o =18 kHz with a frequency divisor of 5 (i.e., the harmonic frequency to which the tank circuit is tuned has a period 1/5 of the period between successive pulses I). Following the conduction period of SCR₁ during which the negative current pulse I occurs, SCR₁ is back-biased by the capacitor voltage e c (t) added to the voltage coupled back from the tank resonator. When SCR₂ conducts to generate the positive current pulse II, forward voltage is re-applied to SCR₁. For Fig. 2, the resonator coupling was set close to the critical level where the back bias applied to SCR₁ just touches zero at the midpoint of the time available for SCR₁ to recover. Any more coupled voltage would cut the back-bias time of SCR₁ in half, and could cause SCR₁ to turn on accidentally, resulting in a collapse of circuit operation. Generically stated, it has been found that if the tank frequency is tuned such that its period T tank = T N , where T is the period between successive similar polarity half wave pulses I, and N is an odd integer greater than 1 (say 3 or 5, for example) the sequentially operated inverter switches SCR₁ and SCR₂ will excite a tank current in successive build-up at such harmonic (Fig. 3), with the coupling inductor L c causing a 90° phase shift between such generated harmonic current in the tank circuit and that harmonic in the pulse currents applied to excite the tank circuit. This is as contrasted from prior art excitation circuits such as in said U.S. Patent 3,786,334 where the period of the frequency at which the tank circuit is tuned to resonate is equal to that of the excitation pulse frequency, requiring a large number of sequential pulses to complete the desired radio-frequency wave or a.c. pulse to be generated. The invention, to the contrary, enables the use of large pulses (I, II), widely spaced and not contiguous, with the period of the tank circuit frequency tuned to be I/N that of the excitation pulse period. As before stated, this is particularly suited to the SCR characteristics of facilely generating short, large current pulses, relatively infrequently. The excitation pulse repetition rate is selected so that the excitation energy of each pulse ringing the tank resonator circuit substantially equals and compensates for the energy dissipated in the ringing oscillations in the tank circuit that was caused by the preceding excitation pulse -- establishing equilibrium in the remainder of the resulting radio-frequency pulse envelope The width adjustment of each half-cycle pulse I or II, together with the amplitude and the harmonic number N, will determine the amount of the harmonically generated current in the tank circuit, Fig. 3A showing the steady-state operation (at, for example, 80 kw on 30.35 kHz). It may be noted that in this equilibrium condition, distortion has appeared in the drive pulse, relative to I and II in the build-up interval of Fig. 3B, demonstrating how a tank or resonator frequency twice the pulse frequency is generated. Fig. 4 is a plot of the harmonic current I N ( ωt ωp ) at angular frequency ω tank contained in a stream of bipolar, one-ampere, half-simesoidal pulses of base width π/ω p seconds and occurring at an angular rate of ω tank /N; i.e., at a rate of one pulse repetition cucle for every N tank-frequency cycles. When multiplied by 4 πN , the vertical axis represents the amps of harmonic current per amp of pulse current, plotted against the ratio of tank frequency ω tank to pulse-width frequency ω pulse. This plot, which is the Nth coefficient in a Fourier cosine series expansion of the pulse stream, gives a first approximation to the harmonic current which a pulse generator can deliver to a load resonant at ω tank . Once there is energy at ω tank circulating in the load tank, a voltage is developed across the tank coupling impedance which opposes the passage of harmonic current, tending to reduce its magnitude below the calculated value I N ( ωt ωp ) and producing a distortion of the basic pulse. This distortion is evident in Fig. 3A, taken with a ratio ω tank /ω pulse of about 2.1. Ratios of 0.7 and 2.1 have been marked on the axis ω tank /ω pulse to show a symmetrical tank-frequency range of 3:1 within which I N ( ωt ωp )≧0.61. By calculation, the generator can be operated over a 10 to 30 kHz range if the basic pulse frequency is set to 14.3 kHz. The absolute maximum tank frequency that could be generated would be 3x14.3=42.9 kHz. In accordance with the invention, the harmonic current amplitude in the tank circuit is controlled by the settings of pulse current amplitude, pulse width and harmonic divisor "N". Especially, the invention provides the ability to operate over a band of tank resonant frequencies with a single generator pulse width simply by changing the pulse repetition rate and possibly the basic divisor "N". Whether the generator pulse width matches, overlaps or underlaps, the resonator half-cycle does not matter materially, as indicated in the graph of Fig. 4. It would be observed that the resonant tank circuit may be considered to be included as part of the generator, rather than as the load, in which event there is provided a sine-wave generator that can drive any sort of load, even another resonant load, such as a VLF antenna. Use of more than one resonator tank will, moreover, improve the quality of the sine-wave delivered to the load, by reducing the sideband content at the pulse-repetition frequency harmonics of order N-1 and N+1. The lower the loaded Q of the resonator, the more noticeable the sidebands will be. Early in the fill-up interval, discharge energy from the d.c. supply enters the pulse generator circuit G much faster than harmonic energy gets delivered to the tank resonator. If left unrestrained, the capacitor voltage e c (t) would build up to destructive levels after just a few discharges. The clamping circuit, so labelled in Fig. 1, establishes a voltage limit for the (symmetrical) positive and negative excursions of e c (t), beyond which some of the discharge current is fed back into the power supply. The clamping level is set at a multiple of the half-supply voltage by a low-leakage autotransformer T₂ and clamping diodes D₁ and D₂. When e c (t) passes the positive threshhold during a discharge through SCR₂ and L₂, diode D₂ conducts and diverts some of the pulse current through T₂ and into the upper d.c. supply. The current out of e c returns to the discharge loop via L₄ to C₂, and via the lower d.c. supply and L₃ to C₁. Fig. 5 shows, for an 18 kHz current circulating in the tank resonator, the current passing from point e c into the clamping transformer. The latter current has proved very useful as a tuning indicator. Limiting the capacitor voltage also limits the pulse discharge current through the main SCRs, so that for load impedances down to a short circuit, the clamped pulse generator behaves like a current source. The pulse-generator circuit of Fig. 1 is also vulnerable to a fault condition called "latch-up" or "shoot-through" wherein both SCR₁ and SCR₂ conduct simultaneously, shorting out the main d.c. supply. Latch-up can be precipitated by false triggers generated in the gating logic, transients which cause forward breakover of a non-conducting SCR, degradation of SCR turnoff characteristics from excessive junction temperature, etc. Current-limiting fuses in the d.c. supply lines can be sized to open before the SCRs are damaged, and provide a useful final defense against latch-up. The nuisance and expense of fuse replacement can be considerable, particularly during development of a new circuit. The circuit in Fig. 1 employs a very effective latch-up inhibitor as explained in U.S. Letters Patent No. 3,832,573. The SCR pulse generator is isolated from the d.c. power supply by coupled coils L₃ and L₄, whose total series inductance, including the additive mutual inductance, is several times that of the discharge inductors L₁ and L₂. Simultaneous conduction of SCR₁ and SCR₂ causes the large filter capacitor C₃ to resonantly discharge through L₁ and L₂, and initiates a recharge of C₃ from the power supply via L₃ and L₄. If losses are low and L₃+L₄≧5(L₁+L₂), the discharge current through SCR₁ and SCR₂ will reverse direction and clear the fault. Capacitor C₃ then resonantly recharges from a slightly negative polarity toward twice the normal supply voltage, a potentially dangerous level. The coupled coils L₃+L₄ are provided with respective companion windings L 1 3 and L 1 4 , as shown in Fig. 1, to provide a voltage step-up ratio 'η' with respect to the total voltage applied to the coupled coil pair L₃+L₄. A diode D₃ in series with the step-up winding is connected so that during normal operation, a back bias of E dc is applied. Following a latch-up, as the recharge of C₃ proceeds beyond the usual operating level of E dc to a higher level E dc +Δ, a voltage ηΔ appears across the step-up winding. Diode D₃ will conduct when ηΔ exceeds E dc , transferring the capacitor re-charging current from L₃+L₄ to the step-up winding and thereby terminating the re-charge of C₃. The current in the step-up winding passes into the power supply, and soon ramps to zero. A voltage-sensing circuit across C₃ detects when a latch-up occurs, and inhibits the SCR gates until the fault clears and the re-charge transient dies out. There is a conflict in the circuit of Fig. 1 between the clamping circuit T₂, D₁ and D₂ and the latch-up protection system. For best clamping, we would like to minimize the circuit inductance in the path e c , T₂, D₂, L₄ around upper capacitor C₂, and in the companion path to the lower side of C₁ via the lower d.c. supply and L₃. But for the latch-up protection circuit to work, we need L₃+L₄≧5(L₁+L₂), as before stated, by mutually coupling the coils L₃ and L₄ in the sense indicated by the dots in Fig. 1. We get the extra mutual inductance 2M₃₄ added to the self-inductance L₃+L₄ for the series discharge circuit mode utilized in latch-up. Better still, for clamping, the (equal) return currents to (equal) capacitors C₁ and C₂ through L₄ and L₃ respectively passes through the coupled assembly in the same direction, so that the (undesireable) total inductance added to the clamping loop is reduced by the mutual coupling to L₃L₄-M 2 34 L₃+L₄+2M₃₄ . For the usual case where the self-inductances L₃ and L₄ are equal, it turns out that the effective "latch-up" inductance is many times the "clamping" inductance of the mutual coupling.

Claims: Apparatus for generating radio frequency pulses or continuous waves comprising a power generator for generating sequential half-cycle pulses of alternately opposite polarity and applying the same to a resonant load, said power generator having first and second saturable inductor switching controls each comprising an SCR (SCR1, SCR2) and a saturable inductor (L1, L2) connected in series, the saturable inductors being connected together to form a common node, a filter capacitor (C3) interconnecting the SCRs (SCR1, SCR2), two series-connected pulse capacitors (C1, C2), being connected in parallel with the filter capacitor (C3), an output transformer (T1) having a primary winding connected between the junction of the said pulse capacitors (C1, C2) and the said common node of the saturable inductors (L1, L2), the secondary winding of the output transformer (T1) being connected to the resonant load, said load comprising a tank resonator circuit provided with tuning means (L, Lc) for tuning the frequency of said tank resonator circuit so that the tank resonator circuit oscillates at a predetermined frequency upon excitation by said power generator, said predetermined frequency being a multiple of the repetition frequency of said half-cycle pulses, a centre-tapped DC supply source (Edc), first and second inductors (L3, L4) connected between the positive terminal of the DC supply source (Edc) and the second SCR (SCR2) and between the negative terminal of the DC supply source (Edc) and the first SCR (SCR1), respectively, first and second series-connected switching diodes (D1, D2) connected across the DC supply source (Edc), with the junction between the switching diodes (D1, D2) connected to the centre tap of an autotransformer (T2) arranged in series between the junction of the said pulse capacitors (C1, C2) and the said centre tap of the DC supply source (Edc) so that when the voltage appearing at the junction between the pulse capacitors (C1, C2) is above a predetermined positive threshold, the second switching diode (D2) conducts to divert the current from the autotransformer (T2) to the DC supply source (Edc), first and second step-up windings (L3′, L4′), each being magnetically coupled to a respective one of said first and second inductors (L3, L4), said step-up windings (L3′,L4′) being connected in series to each other and to a third diode (D3), said series-connected arrangement being connected across the DC supply source (Edc), said step-up windings (L3′, L4′) providing a voltage step-up ratio with respect to the total voltage appearing across the first and second inductors (L3, L4), said third diode (D3) being conductive when the voltage across the filter capacitor (C3) exceeds a predetermined threshold, thereby diverting the current from the first and second inductors (L3, L4) to the corresponding step-up windings (L3′, L4′) and thus terminating the charging of the filter capacitor (C3), so as to prevent simultaneous conduction of the SCRs (SCR1, SCR2) from shorting out the DC supply source (Edc), and a voltage sensing circuit connected across said filter capacitor (C3) for inhibiting said SCRs (SCR1, SCR2) in response to the existence of said excess voltage across the filter capacitor (C3).

Inventors: VERPLANCK PETER, VERPLANCK, PETER
Applicants: MEGAPULSE INC, MEGAPULSE INCORPORATED
Application year: 1987
Main classification: H03K3
IPC codes: H03B11, H03K3