AN AMPLIFIER CIRCUIT SUITABLE FOR USE AS AN ACTIVE FILTER CIRCUIT

Abstract: An amplifier suitable for use as an active filter such as a low-pass filter, high-pass filter or phase shifter, which can be easily formed as part of an integrated circuit. An amplifier comprises first and second transistor (Q₁ and Q₂), which together form a first differential amplifier for convert­ ing the voltage of the input signal thereto to a signal current, third and fourth transistors (Q₃ and Q₄), which together form a second differential amplifier for dividing the signal current from the second transistor (Q₂) in a given ratio, a capacitor (C) connected to the collector of the fourth transistor (Q₄), means connected to the base of the third transistor (Q₃) or fourth transistor (Q₄), said means responsive to a control voltage V c thereto to control the current division ratio (K) for the third and fourth transistors (Q₃ and Q₄), and first and second resistors (R₁ and R₂) connected, respectively, to emitters of the first and second transistors (Q₁ and Q₂). The controlled signal current flows through the load capacitor (C) so that the time constant of an active filter is controlled, The control voltage V c also compensates for the inaccuracy of the values of the resistors and capacitors. Resistors (R₁ and R₂) connected to the emitters of the first and second transistors (Q₁ and Q₂) function to provide the first and second transistors (Q₁ and Q₂) with greater dynamic ranges against the input signal, and also allow them to operate at emitter current with an increasing transition frequency. This improves the high frequency characteristics of the first differ­ ential amplifier.

Description: This invention relates to an amplifier circuit, and particularly to an amplifier circuit suitable for use as an active filter circuit, which can be easily formed into an integrated circuit (IC). When an attempt is made to allow an IC (integrated circuit) to comprise such circuits including filters as integrator and phase shift circuits, it is important how these filters should be formed in the IC so as to decrease the number of components to be externally connected. As filters for use in an integrated circuit, active filters are usually used. However, disadvantages of a conventional active filter are: (1) it is difficult to form its resistors and capacitors with accurate values, causing errors in its cut-off frequency which is determined by the product of these values; and (2) it is difficult to form a circuit with a lower cut-off frequency because resistors and capacitors are limited to have relatively low values. Conventional active filters are disclosed in JP-A-55-45224, JP-A-55-45266 and DE-A-2645632. The DE-A-2645632 discloses an amplifier which comprises first and second transistors which work as an first differential amplifier, a third and fourth transistor, which work as a second differential amplifier. This second differential amplifier is connected to the collector of one of the first two transistors. The collector of one transistor of the second differential amplifier is connected with the output terminal. The two differential amplifiers of the DE-A-2645632 are, however, part of an automatic gain control. An amplifier according to the present invention is defined in Claim 1. It comprises first and second transistors Q₁ and Q₂, which together form a first differential amplifier for converting the voltage of the input signal thereto to a signal current, third and fourth transistors Q₃ and Q₄, which together from a second differential amplifier for driving the signal current from the second transistor Q₂ at a given division ratio, an output terminal of said second differential amplifier circuit being connected to the collector of said fourth transistor Q₄, a capacitor C connected to the collector of the fourth transistor Q₄, means connected to the base of the third transistor Q₃ or fourth transistor Q₄, said means responsive to a control voltage V c thereto to control the current division ratio for the currents of the third and fourth transistors Q₃ and Q₄, and first and second resistors R₁ and R₂ connected, respectively, to emitters of the first and second transistors Q₁ and Q₂. In the amplifier according to the present invention, the signal current shunted in the current division ratio K by the third and fourth transistors Q₃ and Q₄ flows through the load capacitor C so that the time constant of the active filter is controlled, the ratio K being controlled by the control voltage V c . The control voltage V c also compensates for the inaccuracy of the values of the resistors and capacitors. The resistors R₁ and R₂ connected to the emitters of the first and second transistors Q₁ and Q₂ function to provide the first and second transistors Q₁ and Q₂ with greater dynamic ranges against the input signal, and also to allow the first and second transistors Q₁ and Q₂ forming the first differential amplifier to operate constantly at emitter current with increasing transition frequency. This improves the high frequency characteristics of the first differential amplifier. Figs. 1 and 7 are schematic diagrams of amplifiers according to the present invention, Figs. 2 and 8 are schematic diagrams of amplifiers according to the present invention, which are intended for use as integrators, Fig. 3 is a diagram illustrating a relationship between the transition frequency and emitter current of a transistor, Figs. 4 and 9 are schematic diagrams of amplifiers according to the present invention, which are intended for use as low-pass filters, Figs. 5 and 10 are schematic diagrams of amplifiers according to the present invention, which are intended for use as high-pass filters, Figs. 6 and 11 are schematic diagrams of amplifiers according to the present invention, which are intended for use as phase shifters, Figs. 12 and 13 are schematic diagrams of circuits utilizing the low-pass filters shown in Figs. 4 and 9, and Fig. 14 is a block diagram illustrating a typical filter. DESCRIPTION OF THE PREFERRED EMBODIMENTS Fig. 1 shows a schematic diagram illustrating an amplifier according to the present invention. Referring to Fig. 1, a differential amplifier is composed of a transistor Q₁ connected to one end of a resistor R₁ at its emitter and a transistor Q₂ connected to one end of a resistor R₂ at its emitter, the other ends of the resistors being connected to a first regulated current source A₁. The base of the transistor Q₁ is connected to one input T₁, the base of the transistor Q₂ is connected to another input T₂, and an input signal V in is provided across the inputs T₁ and T₂. The collector current of the transistor Q₂ is shunted by transistors Q₃ and Q₄, the emitters of which are connected to the collector of the transistor Q₂. The transistors Q₃ and Q₄ together form a differential amplifier. The base of the transistor Q₃ is supplied with a bias viltage V B1 through a terminal T₈, while the base of the transistor Q₄ is connected to a terminal T₃. The current division ratio in which the current is divided between the transistors Q₃ and Q₄ is controlled by a control voltage V c applied to the terminal T₃. The collector of the transistor Q₄ is connected both to the collector of a transistor Q₅ and to a capacitor C. The collector of the transistor Q₄ is also connected to a terminal T₄ at which an output voltage V out is produced. The capacitor C is connected to a terminal T₅. The base of the transistor Q₅ is connected to the base and collector of a transistor Q₆, and the transistors Q₅ and Q₆ thus form a current mirror circuit in which the transistors Q₅ and Q₆ produce substantially the same collector current. Transistors Q₇ and Q₈ are connected together at their emitters which are connected to a second regulated current source A₂. The current from the regulated current source A₂ is selected to be half the current I₀ from the regulated current source A₁, that is, I₀/2. The bases of the transistors Q₇ and Q₈ are connected to the bases of the transistors Q₃ and Q₄, respectively, and the current division ratio between the transistors Q₃ and Q₄ is controlled by a control voltage V c applied to the terminal T₃. Therefore, the collector currents of the transistors Q₄ and Q₈ are substantially equal, and the collector currents of the transistors Q₃ and Q₇ are substantially equal. Fig. 2, in which like reference numerals represent like elements as in Fig. 1, shows a circuit wherein another bias voltage V B2 is applied to the terminal T₂ with the terminals T₅ and T₆ being connected together. When an input signal V in is supplied to the terminal T₁ in Fig. 2, the collector current i s of the transistor Q₂ driven by the input signal V in is represented by: wherein r e is the emitter resistance of each of the transistors Q₁ and Q₂. Let K be the current division ratio (the portion for the collector of Q₄) of the current i s between the transistors Q₃ and Q₄, then a current Ki s flows through the capacitor C and produces an output voltage V out at the terminal T₄. where S is Raplacean, and ω is the angular frequency of the signal. From the formulas (1) and (2) This indicates that the circuit shown in Fig. 2 is an integrator, the gain of which is given by K/C·(R₁+R₂+2r e ) . The emitter resistance r e is a function of the emitter current I E and given by where k is Boltzman's costant, T is absolute temperature, and q is electron charge. The emitter resistance r E =260Ω at the emitter current of 100 µA is obtained at an amibient (room) temperature. The transition frequency f T , which generally indicates the high frequency performance of a transistor, varies according to the emitter current I E as shown in Fig. 3. When the emitter current I E is selected relatively large, the transition frequency f T increases and thus the high frequency performance improves. If the integral gain is determined by the emitter resistor r e , the transition frequency f T decreases. According to the present invention, resistors R₁ and R₂ and transistors Q₃ and Q₄ are provided to allow transistors Q₁ and Q₂ to operate constantly at the emitter currents I E with increasing transition frequency f T . If the values of the resistors R₁ and R₂ are both selected to be a resistance value R E which is sufficiently large compared to the emitter resistance r e , the formula (3) may be practically rewritten as follows: which means that the integral gain is determined by the resistors R₁ and R₂ and the current division ratio K. If there is a change in the integral gain due to inaccuracy of the value of the capacitor C or resistor R₁, then the control voltage V c supplied to the control terminal T₁ is controlled to control the current division ratio K for the signal current to the collector of the transistor Q₄, so that a desired integral gain can be obtained. Further, the resistance R E serves to provide a greater dynamic range for the input signal, thereby providing the circuit with improved characteristics against distortion and noise. Since the current division ratio K also varies with the bias voltage V B1 supplied to the terminal T₈, the integral gain may be controlled by the voltage V B1 supplied to the terminal T₈. Fig. 4 shows an amplifier according to another embodiment of the present invention, which is intended for use as an integrator. Transistors Q₁₀, Q₁₁, Q₁₂ and Q₁₃ and a regulated current source A₃ form a DC level shift emitter follower. The integral output from a capacitor C is coupled through a terminal T₄ to the base of the transistor Q₁₀, which in turn produces at its emitter an output signal V out which appears at a terminal T₉. The output signal V out is also supplied through the transistors Q₁₁, Q₁₂ and Q₁₃ to a terminal T₂ connected to the base of a transistor Q₂, which together form a negative feedback loop. Therefore, the transfer function H₂(S) of the circuit shown in Fig. 4 is determined by Thus, the circuit shown in Fig. 4 is a low-pass filter with a cut-off frequency ω c given by In this circuit, the inaccuracy of the capacitor C and resistor R E is perfectly canceled by the current division ratio K being controlled. If the values of the capacitor C and resistor R E are relatively small, the current division ratio K is selected to be smaller accordingly so that the cut-off frequency ω c becomes low. It was difficult to form an IC (integrated circuit) which includes a filter with a relatively low cut-off frequency. According to this invention, however, such an IC can be easily formed by utilizing the integrator having a low value capacitor C and low value resistors R₁ and R₂ as shown in Figs. 2 and 4, because the integrator can operate as a filter with a low cut-off frequency. Fig. 5 shows an amplifier according to an embodiment of the present invention, which is intended for use as a high-pass filter. An input signal V in is supplied through a terminal T₅ to one end of a load capacitor C, the other end of which is connected to the collector of a transistor Q₄. The base of the transistor Q₁ is connected to the terminal T₁ connected to a bias voltage V B3 . The transfer function H₃(S) is determined by Thus, the circuit shown in Fig. 5 functions as a high-pass filter. Fig. 6 shows an amplifier according to an embodiment of the present invention, which is intended for use as a phase shifter. In this circuit, the collector of the transistor Q₁ is connected to a load resistor R₃ and the base of a transistor Q₉. A regulated current source A₄ is connected to the emitter of the transistor Q₉, which thus forms an emitter follower. A terminal T₅ connected to the capacitor C is connected to the emitter of the transistor Q₉. The transistor Q₁₃ has an emitter connected to the base of the transistor Q₂ through the terminal T₂ and also to a terminal T₁₀ at which an output signal V out appears. The current i s through the collector of the transistor Q₂ is given by When the current i s flows through the capacitor C, a voltage appears across the capacitor C. Simultaneously, a current equal in amount and opposite in phase to the current i s flows through the collector of the transistor Q₁ to the load resistor R₃. This current produces a voltage across the resistor R₃, and this voltage is then added to the voltage across the capacitor C through the transistor Q₉ and capacitor C. The added signal passes through the transistors Q₁₀, Q₁₁, Q₁₃ and Q₁₄. Thereafter, this signal turns out to be an output signal V out at a terminal T₁₀, and is simultaneously supplied to the base of the transistor Q₂ so that a negative feedback loop is formed. The output signal V out is represented by where i s is the angular frequency of the signal. From the formulas (9) and (10), the transfer function H₄(S) of this circuit is In this formula, the term R₃/(2r e +R₁+R e ) represents the gain of the signal appearing as the voltage across the load resistor R₃. If the term is selected to be 0.5, then the denominator and numerator of the formula are equal, that is, the circuit gain is 1 regardless of the input signal frequency, and the circuit shown in Fig. 6 functions as a phase shifter in which only the phase of the output signal varies with a change in frequency. If the values of the resistors R₁ and R₂ are selected to be sufficiently larger than the emitter resistance value r e so that the gain of the signal across the resistor R₃ will be determined by the resistors R₁, R₂ and R₃, and if the values of the resistors R₁ and R₂ are selected to be equal to the value of R E , then the formula (11) is substantially equivalent to If this phase shifter is formed with inaccurate capacitance C and resistance R E in an IC, the control voltage V c supplied to the terminal T₃ is controlled to control the current division ratio K for the signal current through the transistor Q₄, so that the value of K/(C2R E ) is maintained to be constant. In a single IC (integrated circuit) chip including a plurality of resistors, ratios between their resistance values are substantially equal, so that the gain R₃(2R E ) for the voltage signal across the resistor R₃ remains substantially constant. According to this invention, therefore, the probable inaccuracy of the capacitance and resistance composing the circuit of Fig. 6 in an integrated circuit is compensated for by controlling the current division ratio K between the transistors Q₃ and Q₄, and the current value from the regulated current source A₁ can be selected to a desired value. It is thus possible to select the current value of the regulated current source A₁ for an increasing transition frequency f T of the transistors Q₁ and Q₂, as shown in Fig. 3. This results in improved characteristics in high frequency regions of the transistors Q₁ and Q₂. In the circuit shown in Fig. 1, the output resistance r c of the transistors Q₄ and Q₅ on the side of the terminal T₄ would be infinitive if they were ideal current source. However, the output resistance r c of the transistors Q₄ and Q₅ actually decrease with the increasing collector currents. Especially when formed in an IC chip, the NPN transistor Q₄ can have a greater output resistance, but the PNP transistor Q₅ can have only a smaller output resistance up to several tens or hundreds of kΩ. This will cause the output signal V out at the terminal T₄ to attenuate in the frequency range lower than the frequency determined by the product of the output resistance r c and the load capacitance C, resulting sometimes in such a problem as the inability of the circuit to operate properly. Circuits as will be described may be used to avoid this problem. Fig. 7 shows an improved circuit of the amplifier circuit shown in Fig. 1 according to this invention. Referring to Fig. 7, a transistor Q₂₁ is connected to the transistor Q₅ to form a cascode arrangement. That is, the collector of the transistor Q₄ is connected to the collector of the transistor Q₂₁. A bias voltage V B4 is connected to the base of the transistor Q₂₁, the emitter of which is connected to the collector of the transistor Q₅, both transistors providing a cascode arrangement. The base of the transistor Q₅ is connected to the base and collector of the transistor Q₆ so that a current mirror circuit is formed. The collector current of the transistor Q₅, which is substantially equal to the collector current of the transistor Q₆, passes through the collector of the transistor Q₂₁ as a substantially regurated current. Because of the cascode arrangement of the transistors Q₅ and Q₂₁, the impedance Zo of the transistor Q₂₁ at the terminal T₄ is practically represented by Zo = h fe r c where r c and h fe are the output resistance and current amplification factor of the transistor Q₂₁, respectively. Thus, by connecting the PNP transistor Q₂₁ in a cascode arrangement as a load current source, the output impedance is increased to h fe times the output resistance r e of the transistor Q₂₁. Fig. 8 shows an improved circuit of the integrator shown in Fig. 2. This circuit also comprises a cascode arrangement of the transistors Q₅ and Q₂₁, and the output impedance is increased to h fe times the output resistance r e of the transistor Q₂₁ by connecting the PNP transistor Q₂₁ in a cascode arrangement as a load current source. Therefore, if the output resistance of the transistor Q₄ is sufficiently larger than that of the transistor Q₂₁, the operating range in low frequencies of the integrator also becomes h fe times greater. Fig. 9 shows an improved circuit of the low-pass filter shown in Fig. 4. This circuit also comprises a cascode arrangement of the transistors Q₅ and Q₂₁, and the output impedance is increased to h fe times the output resistance r e of the transistor Q₂₁. Fig. 10 shows an improved circuit of the high-pass filter shown in Fig. 5. This circuit also comprises a cascode arrangement of the transistors Q₅ and Q₂₁. Fig. 11 shows an improved circuit of the phase shifter shown in Fig. 6. This circuit also comprises a cascode arrangement of the transistors Q₅ and Q₂₁. Figs. 12 and 13 show examples of the application of the low-pass filters shown in Figs. 4 and 9. Referring to Fig. 12, the circuit comprises two units of the circuit shown in Fig. 4. In Fig. 13, the circuit comprises two units of the circuit shown in Fig. 9. In Fig. 12, a first low-pass filter is composed of transistors Q₁₀₁, Q₁₀₂, Q₁₀₃, Q₁₀₄, Q₁₀₅, Q₁₁₀, Q₁₁₁, Q₁₁₂ and Q₁₁₃, current sources A₁₀₁ and A₁₀₃, resistors R₁₀₁ and R₁₀₂, and a capacitor C₁₀₀, while a second low-pass filter is composed of transistors Q₂₀₁, Q₂₀₂, Q₂₀₃, Q₂₀₄, Q₂₀₅, Q₂₁₀, Q₂₁₁, Q₂₁₂, and Q₂₁₃, current sources A₂₀₁ and A₂₀₃, resistors R₂₀₁ and R₂₀₂, and a capacitor C₂₀₀. In Fig. 13, additional transistors Q₁₂₁ and Q₂₂₁ are provided in the first and second low-pass filters, respectively. In Figs. 12 and 13, the current division ratio K between the transistors Q₁₀₃ and Q₁₀₄ and between the transistors Q₂₀₃ and Q₂₀₄ is controlled by the control voltage applied to the terminal T₈, and a bias voltage V B5 is applied to the terminal T₃. Fig. 4 shows a typical block diagram of the circuits shown in Figs. 12 and 13. Referring to Fig. 14, an input terminal T₁₁ is connected to index circuits 51, 52 and 53, the outputs of which are connected to arithmetic circuits 54, 55 and 56, respectively. An integrator 100 is connected between the arithmetic circuits 54 and 55, and an integrator 200 is connected between the arithmetic circuits 55 and 56. The arithmetic circuits 54, 55 and 56 are connected to an output terminal T₁₂. The transfer function H₅(S) of the circuit shown in Fig. 14 is given by where G₁ and G₂ are integral gains of the integrators 100 and 200, respectively, and c, b and a are factors of the index circuits 51, 52 and 53. The circuit shown in Fig. 14 functions as a low-pass, high-pass or bandpass filter according to the values of the factors a, b and c. In the circuits shown in Figs. 12 and 13, the integrators 100 and 200 comprise the first and second low-pass filters, respectively. The factors fo the index circuits 51, 52 and 53 are selected to be a=0, b=0 and c=1. Therefore, the transfer function of the circuits shown in Figs. 12 and 13 is given by which is derived from the formula (13). the circuits shown in Figs. 12 and 13 have low-pass characteristics of the second order. In the folmula (14), R E1 is the resistance of the resistors R₁₀₁ and R₁₀₂, and R E2 is the resistance of the resistors R₂₀₁ and R₂₀₂. The transistor Q₉ compensates for the base current to assure the current mirror operation of the transistor Q₆. As described above, this invention provides an amplifier suitable for use as an acrive filter such as a low-pass filter, high-pass filter or phase shifter, which can be easily formed as part of an integrated circuit.

Claims: An amplifier circuit in a filter circuit comprising: a first differential amplifier circuit including first and second input terminals (T₁ and T₂), first and second transistors (Q₁ and Q₂), a first resistor (R₁), connected to an emitter of said first transistor (Q₁), a second resistor (R₂) connected to an emitter of said second transistor (Q₂), and a first current source (A₁) connected to said first and second resistors (R₁ and R₂) to pass a current I₀ therethrough; a second differential amplifier circuit including third and fourth transistors (Q₃ and Q₄) to divide a current flowing through said second transistor (Q₂), said third and fourth transistors (Q₃ and Q₄) having their respective emitters connected to the collector of said second transistor (Q₂), and an output terminal of said second differential amplifier circuit being connected to the collector of said fourth transistor (Q₄); a capacitor C having one end connected to the collector of said fourth transistor (Q₄) and another end connected to a terminal (T₅); control voltage input terminals (T₃, T₈) connected to said second differential amplifier to receive a control voltage V c for determining a current division ratio K for the currents through said third and fourth transistors (Q₃ and Q₄); a current mirror circuit including fifth an sixth transistors (Q₅ and Q₆) which are interconnected parallelly both at their emitters and at their bases, said fifth transistor (Q₅) connected at its collector to the collector of said fourth transistor (Q₄), and a third differential amplifier including seventh and eighth transistors (Q₇ and Q₈), a second current source connected to emitters of said seventh and eighth transistors (Q₇ and Q₈) for providing a current of approximately I₀/2, wherein the collectors of said eighth and sixth transistors (Q₈ and Q₆) are interconnected, the bases of said seventh and third transistors (Q₇ and Q₃) are interconnected, and the bases of said eighth and fourth transistors (Q₈ and Q₄) are interconnected. An amplifier according to Claim 1, wherein an input signal is supplied to said terminal (T₁), a bias voltage (V B2 ) is supplied to said terminal (T₂), and said terminal (T₅) connected to said capacitor (C) is connected to a power source terminal (T₆) or a ground terminal (T₇). An amplifier according to Claim 1, further comprising a negative feedback circuit including, a tenth transistor (Q₁₀) having a base connected to the collector of said fourth transistor (Q₄), and an eleventh transistor (Q₁₁) connected between the emitter of said tenth transistor (Q₁₀) and said second input (T₂) of said first differential amplifier circuit. An amplifier according to Claim 3, wherein said terminal (T₅) connected to said capacitor (C) is connected to said power source terminal (T₆) or said ground terminal (T₇), and an input signal V in is supplied to said input terminal (T₁). An amplifier according to Cliam 3, wherein the input signal V in is supplied to said terminal (T₅) connected to said capacitor (C), and a bias voltage V B3 is applied to said terminal (T₁). An amplifier according to Claim 3, further comprising: a third resistor (R₃) connected between the collector of said first transistor (Q₁) and said power source terminal (T₆), and a ninth transistor (Q₉) having a base connected to the collector of said transistor (Q₁), a collector connected to said power source terminal (T₆), and an emitter connected to said terminal (T₅). An amplifier according to Claim 1 further comprising a transistor (Q₂₁) having an emitter connected to the collector of said fifth transistor (Q₅), a collector connected to the collector of said fourth transistor (Q₄), and a base adapted to receive a bias voltage V B4 .

Inventors: KAMIYA, MASANORI, KUROYANAGI, TOMOMITSU HITACHI ISOGORYO, MATSUMOTO, SHUZO, UEKI, YUKINARI HITACHI MITAKERYO
Applicants: HITACHI, LTD.
Application year: 1987
Main classification: H03G1
IPC codes: H03H11, H03G1