HIGH-ACCURACY POSITION DETECTION APPARATUS

Abstract: A high-accuracy position detection apparatus comprises a position detector (14, 16) for detecting a position signal (ϑB) indicating the rotation position of a motor (10); a coupler (12, 13) for coupling the motor (10) to the position detector (14, 16) with a certain eccentricity; a coupling eccentricity detector (200 - ­ 201), connected to the position detector (14, 16), for detecting an eccentricity signal (Δϑ), corresponding to a coupling eccentricity of the coupler (12, 13), from the position detection signal (ϑB); and a true position signal synthesizer (18), connected to the position detector (14, 16) and the coupling eccentricity detector (200 - 201), for combining the position signal (ϑB) with the eccentricity signal (Δϑ) so as to produce a true position signal (ϑRT) which indicates the true position of the motor (10) and is independent of the coupling eccentricity of the coupler (12, 13).

Description: The present invention relates to a position detection apparatus for detecting the position of an object which is periodically or repetitively moved or the rotation position of a rotary member such as a motor, and a speed detection apparatus to which the position detection apparatus of the invention is applied. An apparatus using a position detector such as an optical encoder or resolver is known, which can detect the rotation position of, for example, a motor. The rotating shaft of the position detector is coupled to a rotating shaft of a motor via a mechanical joint. For this reason, when in use, eccentricity inevitably occurs between these rotating shafts. Any position signal received by the position detector includes a rotating shaft eccentricity signal which at any time is equal to the rotation frequency of the motor and has a given amplitude, due to the eccentricity of the rotating shafts. A differentiated position signal, obtained by differentiating the detected position signal over time, is utilized for controlling the speed of the motor. However, the rotating shaft eccentricity signal cannot be removed by this differentiation process. If speed control is performed using the differentiated position signal, a ripple component corresponding to the eccentricity of the rotating shafts is generated. The ripple component noticeably degrades the performance of the controllability of the motor speed. A configuration of the above apparatus is disclosed in US-A-4,593,193. US-A-4 472 669 discloses a compensation of phase deviations in a resolver detektor. A phase delay in a resolver feedback signal, caused by an analog filter, is compensated for. It is an object of the present invention to provide a position detection apparatus wherein any eccentricity of a coupled rotating shaft of a position detector is compensated for, so that a position detection signal free from the influence of the eccentricity can be obtained, despite any eccentricity of the coupled rotating shaft. It is another object of the present invention to provide a position detection apparatus wherein any eccentricity of a coupled rotating shaft of a position detector is electrically and automatically compensated for, so that a position detection signal free from the influence of the eccentricity of the coupled rotating shaft can be obtained, regardless of the eccentric state thereof. In order to achieve the first object, in a position detection apparatus of the present invention, a deviation (Δϑ) between the position detection signal (ϑA or ϑB), including a ripple component arising from the eccentricity of a coupled rotating shaft, and a predetermined signal (PGT), obtained when there is no eccentricity, is detected. The ripple component in the position detection signal (ϑA or ϑB) is canceled by the deviation (Δϑ), thus enabling a position detection signal free from the influence of the eccentricity of the coupled rotating shaft. To achieve the second object, in a position detection apparatus of the present invention, a linear approximation signal (ytn), obtained by linear-approximating one period (or N periods; N is an integer) of the eccentricity, is detected from a position detection signal (ϑA or ϑB) including a ripple component arising from the eccentricity of a coupled rotating shaft, provided that the speed of the rotating shaft is substantially constant. The linear approximation signal (ytn) is corrected to a predetermined linear signal (PGT) which would be obtained were there no eccentricity of the coupled rotating shaft. A deviation (Δϑ) between the position detection signal (ϑA or ϑB) including the ripple component and the predetermined linear signal (PGT) is detected. The ripple component in the position detection signal (ϑA or ϑB) is cancelled by the deviation (Δϑ), thus enabling a position detection signal free from the influence of the eccentricity of the coupled rotating shaft. This invention can be more fully understood from the following detailed description when taken in conjunction with the accompanying drawings, in which: Fig. 1 is a block diagram showing a basic arrangement of a position detection apparatus according to an embodiment of the present invention; Fig. 2 is a block diagram showing in detail the arrangement shown in Fig. 1; Fig. 3 is a graph for exemplifying a change of one period of a position detection signal (ϑB) over time, obtained by the arrangement shown in Fig. 2; Fig. 4 is a block diagram showing a basic arrangement of a position detection apparatus according to another embodiment of the present invention; Fig. 5 is a waveform chart for explaining the operation of Fig. 4, etc.; Fig. 6 is a block diagram showing a detailed arrangement of Fig. 4; Fig. 7 is a block diagram showing another detailed arrangement of Fig. 4; Fig. 8 is a flow chart showing a sequence for obtaining a deviation (Δϑ) with respect to a predetermined function (PGT or ϑ and) of the position detection signal (ϑB, ϑ) by sampling the position detection signal (ϑB, ϑ) shown in Fig. 3 or 5; Fig. 9 is a block diagram for exemplifying a detailed arrangement of a detector (206, 208) shown in Fig. 6 or a detector (206A, 208A) shown in Fig. 7; and Figs. 10A and 10B are waveform charts for explaining the operation of the arrangement shown in Fig. 9. Embodiments of the present invention will now be described, with reference to the accompanying drawings. For the sake of simplicity, common or similar portions are denoted by common or similar reference numerals in the following description. A position detection apparatus using a resolver as a position detection means will now be described with reference to Fig. 1. In the position detection apparatus, motor 10, as an object to be detected, is coupled via mechanical joint 12 to resolver 14 for detecting the rotation position of motor 10. Detection signal ϑA from resolver 14 is converted by converter 16 to rotation angle signal ϑB. Rotation angle signal ϑB from converter 16 is input to compensator 20, which outputs preset compensation value signal Δϑ which is in one-to-one correspondence with rotation angle signal ϑB. Compensation value signal Δϑ from compensator 20 is added by adder 18 to rotation angle signal SB from converter 16, thereby compensating for rotation angle signal ϑB. In this manner, the true rotation position ϑRT of motor 10 can be obtained from adder 18. Fig. 2 is a block diagram showing in detail the arrangement shown in Fig. 1. In the position detection apparatus shown in Fig. 2, detection signal ϑA from resolver 14 is converted to rotation angle signal ϑB by resolver-output-to-rotation-angle converter (to be referred to as R/T converter) 16. True rotation position deducer 200 simulates linear function PGT, which represents the deduced true rotation position of motor 10, based on rotation angle signal ϑB from R/T converter 16 and detection time t of signal ϑB. Deviation Δϑ between output signal ϑ and from deducer 200 and rotation angle signal ϑB from converter 16 is calculated by compensation value calculator 201. Compensation value signal Δϑ from compensation value calculator 201 is added by adder 18 to rotation angle signal ϑB from converter 16, thereby enabling the true rotation position function ϑRT of motor 10 to be calculated. The operation of the position detection apparatus having the above arrangement will now be described with reference to Fig. 3. First, detection signal ϑA of the rotation position of motor 10, detected by resolver 14 which is mechanically coupled thereto, is converted by converter 16 to rotation angle signal ϑB. Rotation position ytn of motor 10 is simulated in accordance with the following equations, using rotation angle signal ϑB in true rotation position deducer 200, based on time t0, at which the rotation position of motor 10 is at rotation start point PS, rotation start position ϑBS at time t0, time t1, at which the rotation position is at rotation end point PE, and rotation end position ϑBE at time t1: ϑBS = at0 + b (1) ϑBE = at1 + b (2) ytn = {(ϑBE - ϑBS)/(t1 - t0)}·tn + {(t1ϑBS - t0ϑBE)/(t1 - t0)} (3) where ytn is the deduced rotation position, a and b are constants, and tn is the detection time (tn = tn1 to tn9 in Fig. 3). A deviation (ϑtn - ytn) between deduced rotation position ytn at time tn and signal ϑB is obtained by deducer 200, in accordance with equation (3), based on rotation angle signal ϑB obtained from converter 16 during a single rotation of motor 10 (period T) and time tn, at which rotation angle signal ϑB is detected or sampled. The obtained deviation is averaged to form linear equation PGT, which represents a deduced true rotation position. The linear equation is obtained as follows: where ΔP denotes a deviation compensation value, tn denotes the detection time, t0 denotes the time at which the rotation position of motor 10 is present at rotation start point PS, and t1 denotes the time at which the rotation position of motor 10 is present at a rotation end point after a single rotation from rotation start point PS. ϑtn denotes the rotation angle signal (ϑA or ϑB) detected at time tn, ytn denotes the deduced rotation position signal at time tn, and n is the number of detection times (or the number of sampling times of tn). From equations (3) and (4), deduced true rotation position function PGT of motor 10 is simulated in accordance with the following equation: From equation (5), deduced true rotation position function PGT of motor 10 can be calculated by deducer 200. Deducer 200 supplies signal ϑ and corresponding to function PGT to compensation value calculator 201. Calculator 201 calculates deviation Δϑ between output signal ϑ and from deducer 200 and rotation angle signal ϑB from converter 16, thereby deriving a calibrated compensation value: Δϑ= ϑB - ϑ (6) The compensation value representing deviation Δϑ is added by adder 18 to rotation angle signal ϑB from converter 16, whereby true rotation position ϑRT of motor 10 is obtained, free from the influence of the ripple component arising from the eccentricity of joint 12. Conventionally, an error in mounting (joint eccentricity of joint 12) the resolver inevitably occurs due to machining precision, and this prevents any improvement in position detection precision. In the embodiment shown in Fig. 2, a compensation value corresponding to the error is automatically generated and calibrated. As a result, position detection precision, better than machining precision of the joint between the motor and the resolver, can be obtained. In this embodiment, the resolver serves as the position detection means. The present invention can be applied to an optical position detecting apparatus (e.g., an optical encoder). The present invention can also be applied to a speed detector for detecting position signal (ϑA or ϑB) at predetermined time intervals, and for ascertaining the operating speed of an object (such as a motor) on the basis of the amount of the movement of the object during this interval. A speed detector apparatus, to which the position detection apparatus of the present invention is applied, will now be described, with reference to Fig. 4. As is shown in Fig. 4, rotation angle ϑA of motor 10, to be detected, is detected by resolver 14 of the speed detector. Rotating shaft 12A of motor 10 is coaxially fixed to detection shaft 12B of resolver 14 via coupler 13. Compensation angle Δϑ is generated from compensation function generator 202, as a function of output signal ϑA, based on output signal ϑA from resolver 14. Output signal Δϑ from generator 202 is subtracted, by actual position calculator 162, from output signal ϑA from resolver 14, whereby actual rotation angle ϑRT of motor 10 is obtained. Output signal ϑRT from calculator 162 is differentiated by actual speed calculator 22, thereby calculating actual rotation speed ωT of motor 10. (Note that components 162 + 202 in Fig. 4 correspond to components 16 + 18 + 200 + 202 in Fig. 2.) If, in the speed detector with the above arrangement, motor 10 is rotated at a constant speed, rotation angle ϑ (or ϑA) can be detected by resolver 14 as a sine waveform, indicated by a solid curve in Fig. 5. Detected rotation angle ϑ is input into compensation function generator 202 having a compensation table corresponding to rotation angle ϑ, and compensation angle Δϑ is calculated thereby. Calculated compensation angle Δϑ is subtracted from rotation angle ϑA detected by resolver 14, whereby rotation angle ϑRT of motor 10, corresponding to the actual amount of rotation, is obtained. Rotation angle ϑRT is differentiated over time by calculator 22, to thereby produce actual rotation speed ωT of motor 10. When, in the above speed detector, the compensation angle is preset in a fixed compensation table in calculator 202, then if coupler 13 coaxially coupling motor 10 and resolver 14 is detached and thereafter is recoupled, a problem occurs. Specifically, if the mounting position of coupler 13 is changed upon recoupling, a compensation function in the compensation table, which has been determined prior to re-coupling, must then be reset. If not, resultant rotation angle ϑRT includes a ripple component. When a position detector in which angle signal ϑA represents N rotations (N is a natural number) upon single rotation of motor 10 is used, the rotation position of motor 10 cannot be detected upon ON/OFF of the power source of the apparatus shown in Fig. 4, and a compensation function entered into the compensation table cannot be used. Figs. 6 and 7 show position detection apparatuses which are free of the above-mentioned drawbacks. (Although a speed detector free from the above drawbacks can be realized by differentiating output signal ϑRT, shown in Fig. 2 over time, a new embodiment will be described hereinafter.) The position detection apparatus shown in Fig. 6 has resolver 14 for detecting rotation angle ϑA of motor 10, and coupler 13 for coaxially coupling rotating shaft 12A of motor 10 with detection shaft 12B of resolver 14. Output signal ϑA from resolver 14 is differentiated by speed detector 204, and is output as speed signal ωS. Band-pass filter 205 extracts, from output signal ωS, error signal ωM having the same frequency as that of output signal ϑA from resolver 14. Phase difference PD between output signal ϑA from resolver 14 and output signal ωM from band-pass filter 205 is detected by phase difference detector 206. A low-frequency component of output signal PD from detector 206 is extracted by low-pass filter 207. The amplitude of output signal ωM from filter 205 is divided, together with output signal ωS from detector 204, by amplitude detector 208. The low-frequency component of output signal A from detector 208 is extracted by low-pass filter 209. Output signal ϑA from resolver 14, output signal PDL from low-pass filter 207, and output signal AL from low-pass filter 209 are input to sine wave generator 210, thereby calculating error angle signal ϑM included in output signal from ϑA resolver 14. Output signal ϑM from generator 210 is subtracted from output signal ϑA from resolver 14 by actual angle calculator 164, thus calculating actual angle signal ϑRT which has the same frequency as the rotation speed of motor 10 and does not include an angular error. Output signal ϑRT from calculator 164 is differentiated over time by actual speed calculator 22, and as a result, actual speed signal ωT of motor 10 is calculated. In the speed detection apparatus having the above arrangement, the speed of motor 10 is calculated as follows: Angle signal ϑA detected by and output from resolver 14 includes angular error ϑM due to eccentricity of detection shaft 12B to the center of rotating shaft 12A, with respect to actual rotation angle ϑRT of motor 10. Since angular error ϑM is synchronous with angle signal ϑA, the following equation can be established (where ϑ = ϑA, in Fig. 1, ϑ = ϑB can be adopted): ϑ = ϑRT + ϑM = ϑRT + ALsin(ϑ + PDL) (7) where A1: the amplitude of angular error ϑM, and PDL: a phase. When angle signal ϑ (= ϑA) is differentiated over time, speed signal ωS including angular error ϑM can be obtained by the following equation: ωS = dϑ/dt = (dϑRT/dt) + (dϑM/dt) = ωT + ωTAL·cos(ϑ + PDL) (8) Therefore, when only the speed signal (angular error component) of the second term of the right side of equation (8) is detected, speed error ωM can be obtained by the following equation: ωM = ωT·AL·cos(ϑ + PDL) (9) Phase PDL and amplitude AL obtained from equation (9) are constants determined by the eccentricity of detection shaft 12B with respect to rotating shaft 12A, and hence, can be treated as DC signals. When phase difference PD and amplitude A are filtered through low-pass filters 207 and 209, an unnecessary component corresponding to a detection error can be removed from from PDL and AL. Furthermore, PDL and AL can be gradually detected during rotation of motor 10. Phase PDL and amplitude AL filtered through low-pass filters 207 and 209 and angle signal ϑ (= ϑA) are input to sine wave generator 210, so that angular error ϑM is obtained by the following equation: ϑM = AL·sin(ϑ + PDL) (10) Angular error ϑM obtained from equation (10) can be subtracted from angle signal ϑ (= ϑA) as a compensation signal so as to obtain actual angle signal ϑRT of motor 10. Finally, angle signal ϑRT is differentiated over time, and actual speed signal ωT from which an error having the same frequency as that of the rotation speed caused by the eccentricity of detection shaft 12B with respect to rotating shaft 12A is removed can be obtained from the following equation: ωT = dϑT/dt = (d/dt)(ϑ - ϑM) (11) So as to improve the detection precision when motor 10 is rotating at low-speed, and when a resolver is used whose angle signal ϑ represents N rotations per 10 rotations of motor 10, detected angle signal ϑ can be frequency-divided by N, using the following equation, so that actual speed ωT can be detected by means of the above equations (7) to (11): ϑ = N·(ϑRT·ϑM) (12) (where N is an integer equal to or greater than 2) Another embodiment of the present invention, in which frequency division by N is performed, will now be described, with reference to Fig. 7. The speed detection apparatus shown in Fig. 7 employs resolver 14A whose angle signal ϑ (= ϑ1A) represents N rotations per rotation of motor 10. Rotating shaft 12A of motor 10 is coaxially fixed to detection shaft 12B of resolver 14A via coupler 13. Output signal ϑ1A from resolver 14A is frequency-divided by N, by frequency divider 212. Output signal ϑ1A from resolver 14A is differentiated by speed detector 204A, to thereby be converted to speed signal ω1S. Band-pass filter 205A detects, from output signal ω1S, error signal ω1M having the same frequency as that of output signal ϑ1N from frequency divider 212. Phase difference P1D between output signal ω1M from filter 205A and output signal ϑ1N from frequency divider 212 is detected by phase difference detector 206A. The low-frequency component of output signal P1D from detector 206A is extracted by low-pass filter 207A. The amplitude of output signal ω1M from band-pass filter 205A is divided, together with output signal ω1S from detector 204A, by amplitude detector 208A. The low-frequency component of output signal A1 from detector 208A is extracted by low-pass filter 209A. Cosine wave generator 210A generates cosine wave signal C having amplitude A1L, based on output signal A1L from low-pass filter 209A, output signal P1DL from another low-pass filter 207A, and output signal ϑ1N from frequency divider 212. Cosine wave signal C from generator 210A and output signal ω1S from detector 204A are multiplied with one another by multiplier calculator 166. Output signal ωD from calculator 166 is subtracted from output signal ω1S from detector 204A by actual speed calculator 220, whereby actual speed signal ω1T of motor 10 is obtained. In the speed detection apparatus having the above arrangement, speed detection is performed as follows: Angle signal ϑ1A detected by and output from resolver 14A includes angular error ϑM, with respect to actual rotation angle ϑRT of motor 10, as a result of the eccentricity between rotating shaft 12A and detection shaft 12B. Since angular error ϑM is synchronous with angle signal ϑ1A, the following equation can be established: ϑ1A = ϑRT + ϑM = ϑRT + A1L·sin(ϑ1A + P1DL) (13) where AL: the amplitude of angular error ϑM, and P1DL: the phase of the second right term of equation (13). Therefore, when angle signal ϑ1A is differentiated over time, speed signal ω1S including angular error ϑM can be obtained by the following equation: ω1S = dϑ1A/dt = (dϑRT/dt) + (dϑM/dt) = ω1T + ω1T·A1L·cos(ϑ1A + P1DL) (14) Speed error ωD can be calculated from the second term of the right side of equation (14), by means of the following equation: ωD = ω1T·A1L·cos(ϑ1A + P1DL) (15) In this embodiment, ω1S can be used in place of ω1T of the second term of the right side. Since ω1S·A1L·cos(ϑ1A + P1DL) = (ω1T + ωD)·A1L·cos(ϑ1A + P1DL) = (ω1T + ωD)·(ωD/ω1T) = ωD + (ωn²/ω1T) (16) ωD²/ω1T becomes a negligible small value. Therefore, from equations (14) and (16), ω1T = ω1S - ω1S·A1L·cos(ϑ1A + P1DL) = ω1S - ωD (17) More specifically, output signal ωD from multiplier calculator 166 is subtracted from output signal ω1S from detector 204A, so that actual speed ω1T, which does not include speed error ωD, arising from angular error ϑM having the same frequency as that of the rotational speed of motor 10, can be obtained. As has been described above, the speed detection apparatus of the present invention can gradually detect an error, having the same frequency as that of the rotation speed of the motor and caused by the joint eccentricity between the motor and the resolver, as a DC value during rotation of the motor. Therefore, a compensation function need not be set, and the speed of the motor can be detected with high accuracy. The following US Patents can be utilized in the present invention: (1) USP 4,527,120 (July 2, 1985); Kurosawa (2) USP 4,556,885 (December 3, 1985); Kurosawa Fig. 8 is a flow chart showing a sequence for obtaining a compensation angle (deviation Δϑ) in the arrangement shown in Fig. 1 or 4. Since this sequence can be applied to either arrangement of Figs. 1 and 4, angle signals ϑA and ϑB are represented as ϑ in this flow chart. The flow chart will now be described, with reference to Fig. 5. (Note that the sequence of this flow is executed by a conventional microcomputer (not shown). The microcomputer is incorporated in compensator 20, shown in Fig. 1, or in function generator 202, shown in Fig. 4.) When angle signal ϑ crosses zero level at time t0, as shown in Fig. 5, the zero cross timing is detected (ST10, YES). Values ϑ(t) of angle signals ϑ at some sampling timings (tn1 to tn9 in Fig. 3) from time t1 to time t1 are detected, and time interval from time t0 to time t1 is measured (ST11). The detection operation is repeated until another zero cross of angle signal ϑ is detected (ST12, NO). When another zero cross of angle signal ϑ is detected (ST12, YES), sampling time t (tn1 to tn9 in Fig. 3) during time interval T is defined as a parameter for the following calculations (ST13). Then, deduced true rotation position signal ϑ and(t) (corresponding to PGT in Fig. 3) is calculated, based on data in time interval T and sampling times t obtained in steps ST11 and ST13 and maximum value ϑmax of angle signal ϑ(ϑ and(t) = (ϑmax/T).t) in step ST14). Then, deviation Δϑt is calculated from ϑ(t) and ϑ and(t) obtained in steps ST11 and ST14 (Δϑt = ϑ(t) - ϑ and(t) In step ST15. Finally, new compensation angle data Δ ϑ(ϑ(t)) is calculated from predetermined weighting data a1 and a2, deviation Δϑt obtained in step ST15, and old compensation angle data Δ ϑ(ϑ(t)) (a1Δϑ(t)) + a2Δϑt → Δϑ(ϑ(t)) in step ST15). Resultant final compensation angle data Δϑ(ϑ(t) is used as the compensation angle (deviation Δϑ) in the arrangement shown in Fig. 1 or 4. Fig. 9 is a block diagram showing a detailed arrangement of detectors 206 and 208 in Fig. 6 or detectors 206A and 208A in Fig. 7. Figs. 10A and 10B are waveform charts for explaining the operation of the arrangement shown in Fig. 9 (in Figs. 10A and 10B, ϑA and ϑB are represented by ϑ.) In Fig. 9, angle detection signal ϑA from resolver 14 (14A) is input to latch 2061 and adder 2063. Speed error signal ωM (ω1M) from band-pass filter 205 (205A) is input to zero cross sensor 2060 and sample & hold circuit 2080. Zero cross sensor 2060 supplies latch pulse LP to latch 2061 at a timing at which signal ωM crosses the zero level (t10 in Fig. 10A). Thus, a value (90°-PDL in Fig. 10B) of angle detection signal ϑ at time t10 is loaded in latch 2061. Note that PDL indicates a phase difference between a maximum amplitude point of signal ωM and the zero cross point of signal ωM. Time t10 corresponds to a time phase-delayed by 90 degrees from the maximum amplitude point of signal ωM. The value (90°-PDL) of angle detection signal loaded in latch 2061 is input to adder 2062. In adder 2062, the value (90°-PDL) is added to fixed data equivalent to -90 degrees, whereby phase difference signal -PDL is obtained. Phase difference signal -PDL is phase-inverted by inverting amplifier 2064 to be converted to phase difference signal PD (P1D). Phase difference signal -PDL is added to angle detection signal ϑA (ϑ1A) by adder 2063. If signal ϑA (ϑ1A) is represented by ϑ, adder 2063 outputs an angle signal equivalent to ϑ - PDL. The most significant bit (MSB) of the angle signal (ϑ - PDL) is supplied to sample & hold circuit 2080 as a sampling timing pulse. Sample & hold circuit 2080 samples speed error signal ωM (ω1M) at a time phase-advanced from the maximum point of the angle signal (ϑ - PDL) by PDL, and holds the sampling result. In this manner, amplitude signal A (A1) indicating amplitude ωT·AL (Fig. 10A) of speed error signal ωM (ω1M) can be obtained from sample & hold circuit 2080. While latch 2061 latches signal ϑA (ϑ1A) and while sample & hold circuit 2080 holds speed error signal ωM (ω1M), phase difference signal PD (P1D) and amplitude signal A (A1) become constant DC values. According to the above embodiments as described above, when an eccentricity occurs between an object to be detected and a resolver, an error having the same frequency as that of a moving amount of the object can be removed without the need to newly prepare a complicated compensation function. Since the phase and amplitude of the movement error can be gradually detected during the operation of the object, a compensation signal can be generated without the need to reset the compensation function. In this invention, since the compensation value is automatically calculated and is calibrated for every predetermined period (T), a high-accuracy detection result (ϑRT, ωT) can be obtained, from a statistical viewpoint, and the work load of the operator (e.g., manual correction of the compensation value) can be eliminated. If detection signal ϑ has a repetition characteristic as that shown in Fig. 5, the present invention can be used for position or speed detection of an object other than a rotating member such as motor 10, provided that the speed of movement of the object is substantially constant during a prescribed interval (e.g., the interval from position PS to position PE in Fig. 3).

Claims: A high-accuracy position detection apparatus comprising: position detection means (14, 16) for detecting a position signal (ϑ, ϑA, ϑB) indicating a position of an object (10) which is periodically and repetitively moved; coupling means (12, 13) for coupling said object (10) to said position detection means (14, 16) with an eccentricity; coupling eccentricity detection means (20, 200 -210), connected to said position detection means (14, 16), for detecting an eccentricity signal (Δϑ, ϑM), corresponding to an eccentricity resulting from the coupling of said coupling means (12, 13), from the position detection signal (ϑ, ϑA, ϑB); and true position signal synthesizer means (18, 162, 164), connected to said position detection means (14, 16) and said coupling eccentricity detection means (20, 200 - 210), for combining the position signal (ϑ, ϑA, ϑB) with the eccentricity signal (Δϑ, ϑM), to provide a true position signal (ϑRT), which indicates a true position of said object (10) and is independent of the coupling eccentricity of said coupling means (12, 13). An apparatus according to claim 1, characterized in that said coupling eccentricity detection means (20, 200 - 210) includes: linear function signal generation means (200) for detecting a linear-approximation signal (ytn) obtained by linear-approximating a change in the position signal (ϑB) during a repetitive movement period (T) of said object (10), and for converting the linear-approximation signal (ytn) into a linear function signal (PGT or ϑ and), obtained by linear approximation, when the coupling eccentricity is minimum; and deviation detection means (201) for detecting a deviation of the position signal (ϑB) from said position detection means (16), with respect to the linear function signal (PGT or ϑ and), and outputting the deviation signal (Δϑ, ϑM). An apparatus according to claim 2, characterized in that said linear-approximation signal (ytn) is defined by a line connecting a signal level (ϑBS) of the position signal (ϑB) at a start point (PS) of the repetitive movement period (T) of said object (10), and a signal level (ϑBE) of the position signal (ϑB) at an end point (PE) of the repetitive movement period (T). An apparatus according to claim 3, characterized in that a line defined by the linear function signal (PGT) is parallel to the line defined by the linear-approximation signal (ytn) and the distance (ΔP) between these lines is obtained by the following equation: where ΔP represents the distance between two parallel lines (PGT, ytn), n represents the number of same time divisions in the repetitive movement period (T), t0 represents time t at the start point (PS), t1 represents time t at the end point (PE), ϑtn represents the signal level of the position signal (ϑB) at nth time tn, when the repetitive movement period (T) is equally divided by n , and ytn represents the linear-approximation signal. An apparatus according to claim 1, characterized by further comprising: speed detection means (22) for differentiating, with respect to time, the true position signal (ϑRT) from said true position signal synthesizer means (162), and outputting a speed signal (ωT) indicating a repetitive movement speed of said object (10). An apparatus according to claim 5, characterized in that said coupling eccentricity detection means (20, 200 - 210) includes: function generation means (202) for converting a change in the position signal (ϑA), during the repetitive movement period (T) of said object (10), to a predetermined linear function signal (ϑ and), detecting a deviation of the position signal (ϑA) from said position detection means (16), with respect to the predetermined linear function signal (ϑ and), and outputting the eccentricity signal (Δϑ, ϑM). An apparatus according to claim 5, characterized in that said coupling eccentricity detection means (20, 200 - 210) includes: first means (204) for differentiating, with respect to time, a change in the position signal (ϑA) during the repetitive movement period (T) of said object (10), and generating a speed signal (ωS) indicating a repetitive movement speed of said object (10); second means (205), coupled to said first means (204), for extracting a component, resulting from the coupling eccentricity, from the speed signal (ωS), and generating an eccentricity error signal (ωM); third means (206, 207), coupled to said second means (205), for generating a phase difference signal (PDL) indicating a phase difference between the eccentricity error signal (ωM) and the position detection signal (ϑA); fourth means (208, 209), coupled to said second means (205), for generating an amplitude signal (AL) indicating an amplitude of the eccentricity error signal (ωM); and fifth means (210), coupled to said third means (207) and said fourth means (209), for generating an AC signal (ϑM) having the same period (T) as the position signal (ϑA), an amplitude determined by the amplitude signal (AL) and a phase determined by the phase difference signal (PDL). An apparatus according to claim 7, characterized in that said third means (206, 207) includes: latch pulse generation means (2060) for generating a latch pulse (LP) when the eccentricity error signal (ωM) has reached a predetermined signal level (0 level); latch means (2061), connected to said latch pulse generation means (2060), for latching the signal level of the position signal (ϑA) when the latch pulse (LP) is generated, and outputting a first phase signal (90°-PDL); first compensation means (2062), connected to said latch means (2061), for adding a predetermined compensation signal (-90°) to the first phase signal (90°-PDL), and outputting a second phase signal (-PDL); second compensation means (2063), connected to said first compensation means (2062), for adding the second phase signal (-PDL) to the position signal (ϑA), and outputting a third phase signal (ϑ-PDL); and means (2064), connected to said first compensation means (2062), for converting the second phase signal (-PDL) into a signal (PD) corresponding to the phase difference signal (PDL). An apparatus according to claim 8, characterized in that said fourth means (208, 209) includes: sample & hold means (2080), connected to said second compensation means (2063), for sampling the signal level of the eccentricity error signal (ωM) when the third phase signal (ϑ-PDL) corresponds to a specific time (MSB of ϑ-PDL) during the repetitive movement period (T) of said object (10), holding the sampled signal level, and outputting a signal (A) corresponding to the amplitude signal (AL). An apparatus according to claim 5, characterized in that said coupling eccentricity detection means (20, 200 - 210) includes: speed signal generation means (204A) for differentiating, with respect to time, changes in the position signal (ϑ1A) during a plurality of repetitive movement periods (N × T) of said object (10), and generating a speed signal (ω1S) indicating a repetitive movement speed of said object (10); frequency-divided signal generation means (212) for frequency-dividing the position signal (ϑ1A) by the number (N) of the plurality of the respective movement periods, and generating a frequency-divided signal (ϑ1N); eccentricity error signal generation means (205A), coupled to said speed signal generation means (204A) and said frequency-divided signal generation means (212), for extracting a component, resulting from the coupling eccentricity, from the speed signal (ω1S) and the frequency-divided signal (ϑ1N), and generating an eccentricity error signal (ω1M); phase difference signal generation means (206A, 207A), coupled to said eccentricity signal generation means (205A) and said frequency-divided signal generation means (212), for generating a phase difference signal (P1DL) indicating a phase difference between the eccentricity error signal (ω1M) and the frequency-divided signal (ϑ1N); amplitude signal generation means (208A, 209A), coupled to said eccentricity error signal generation means (205A), for generating an amplitude signal (A1L) indicating an amplitude of the eccentricity error signal (ω1M); and AC signal generation means (210A), coupled to said frequency-divided signal generation means (212), said phase difference signal generation means (207A), and said amplitude signal generation means (209A), for generating an AC signal (C) having a period (T) synchronous with the frequency-divided signal (ϑ1N), an amplitude determined by the amplitude signal (A1L), and a phase determined by the phase difference signal (P1DL). An apparatus according to claim 10, characterized by further comprising: means (166, 200), connected to said speed signal generation means (204A) and said AC signal generation means (210A), for generating the speed signal (ωT) in correspondence with a product of the AC signal (C) and the eccentricity error signal (ω1M). An apparatus according to claim 1, wherein said position detection means (14) detects a rotation position of a rotary member (10) and includes signal conversion means (16) for converting a detection signal (ϑA) from said position detection means (14) into an arbitrary conversion signal (ϑB); characterized by calculation deducting means (200) for calculating and deducing a true rotation position of said rotary member (10), with reference to the conversion signal (ϑB) from said signal conversion means (16), and outputting a deduced signal (ϑ and); compensation calculation means (201) for calculating a compensation signal (Δϑ) for compensating for the conversion signal (ϑB) from said signal conversion means (16), based on the deduced signal (ϑ and) from said calculation deducting means (200) and the conversion signal (ϑB) from said signal conversion means (16); and compensation means (18) for combining the compensation signal (Δϑ) from said compensation calculation means (201) with the conversion signal (ϑB) from said signal conversion means (16), and calculating a true rotation position signal (ϑRT) of said rotary member (10). A speed detection apparatus comprising: position detection means (13, 14) for detecting the amount of movement of an object (10); speed calculation means (204 204A) for differentiating, with respect to time, a movement amount signal (ϑA, ϑ1A) supplied from said movement amount detection means (13, 14, 14A), and calculating a speed signal (ωS; ω1S); error detection means (205 205A) for detecting an error component (ωM, ω1M) included in the movement amount signal (ϑA), based on the speed signal (ωS) supplied from said speed calculation means (204, 204A) and the movement amount signal (ϑA); phase difference detection means (206, 206A) for detecting a phase difference between an error signal (ωM, ω1M) supplied from said error detection means (205, 205A) and the movement amount signal (ϑA); first filter means (207, 207A) or filtering a predetermined frequency component of a phase difference signal (PD) supplied from said phase difference detection means (206; 206A); amplitude detection means (208, 208A) for detecting the amplitude of the error signal (ωM, ω1M) supplied from said error detection means (205, 205A); second filter means (209, 209A) for filtering another predetermined frequency component of an amplitude signal (A, A1) supplied from said amplitude detection means (208 208A); waveform generation means (210, 210A) for generating a cosine or sine wave signal (C, ϑM), based on a first filtered signal (PDL), supplied from said first filter means (207, 207A), a second filtered signal (AL, A1L), supplied from said second filter means (209 209A), and the movement amount signal (ϑA) from said movement amount detection means (13, 14); and actual speed detection means (164, 22) for detecting, as the actual speed of said object (10), an actual speed signal (ωT, ω1T) from which the error component (ωM, ω1M) removed, based on the sine wave or cosine wave signal (ϑM, C) from said waveform generation means (210, 210A) and the movement amount signal (ϑA) from said movement amount detection means (13, 14) or the speed signal from the speed calculation means (204, 204A). The speed detection apparatus of claim 13, comprising: frequency-dividing means (212) for frequency-dividing the movement amount signal (ϑ1A) from said movement amount detection means (913, 14A); wherein the error detection (205A) calculates an error component (ω1M) included in the speed signal (ω1S), based on the speed signal (ω1S) from said speed calculation means (204A) and a frequency-divided signal from said frequency-dividing means (212); the phase difference detection means (206A) detects a phase difference between the error signal (ω1M) supplied from said error calculation means (205A) and the frequency-divided signal (ϑ1N) from said frequency-dividing means (212); the waveform generation means (210A) generates a cosine wave signal (C) based on a first filtered signal (P1DL), supplied from said first filter means (207A), a second filtered signal (A1L), supplied from said second filter means (209A), and the frequency-divided signal (ϑ1N) output from said frequency-dividing means (212); and the actual speed calculation means (166, 220) calculates as the actual speed of said object (10), an actual speed signal (ω1T) from which the error component (ω1M) is removed, based on the cosine wave signal (C) from said waveform generation means (210A) and the speed signal (ω1S) from said speed calculation means (204A).

Inventors: KUROIWA, AKIHIKO C/O PATENT DIVISION, NAKAMURA, RITAKA C/O PATENT DIVISION
Applicants: KABUSHIKI KAISHA TOSHIBA
Application year: 1987
Main classification: G01D3
IPC codes: G01D3