METHOD OF PRECHARGING AND PRECHARGE CIRCUIT FOR DYNAMIC CASCODE VOLTAGE SWITCH LOGIC

Abstract: A precharge circuit for a cascode voltage switch (14) in which at the beginning of the precharge phase the output state is memorised and the output (1) is isolated from the precharging points (N₀, N G ). Both the positive and negative ends of the discharge paths are precharged with the gates of the switches in all paths held in their memorised states. Towards the end of precharging, the output (1) is reconnected to the normal precharging point (N e ) so that it goes low. Then the positive and negative precharging points (N₀, N G ) are reconnected for their evaluation configuration.

Description: The invention relates generally to logic circuits. In particular, the invention relates to a method for consistently precharging the nodes of a dynamic logic circuit and a precharge circuit for performing said method. Background A recently developed technology for logic circuits is called alternatively a cascode voltage switch (CVS) or CMOS domino circuit. An early description of this type of circuit appears in a technical article by Krambeck et al., entitled "High-Speed Compact Circuits with CMOS" and appearing in IEEE Journal of Solid-State Circuits, volume SC-17, No.3, June 19, 1982 at pp 614-619. A similar circuit is disclosed by Booher in US-A-3,601,627. The overall structure of a generalised but exemplary CVS circuit is shown in Figure 1. An input register 10 provides a plurality of primary inputs of which in this example eight PI₁-PI₈, are explicitly shown. There are generally a plurality of logic modules f₁-f₇, each consisting of a group of logic transistors, having one or more inputs X i and followed by an inverter. The inputs X i of each logic group f are either taken from the primary inputs PI₁ - PI₈ or from the output of the inverter of a logic group f of a previous stage. Selected outputs, three Y₁ - Y₃ are shown in the example, are led to an output register 12. The complemented output of a logic group f and the following inverter follow from the usual implementations of the logic group f. A relatively simple logic group 14 is shown in Figure 2 and is implemented with five n-channel field effect transistors 16 each having a gate input X₁-X₅. The function implemented is -OR(AND(X₁, X₃, X₅), AND (X₂, X₄)). This logic group 14 is presented for sake of example and much more complicated logic groups are possible. A signal node N₀ of the logic group 14 is connected to a power supply (not shown) through an upper precharge p-channel transistor 18. A grounding node N g of the logic group 14 is connected to ground potential (not shown) through a lower precharge n-channel transistor 20. In a precharge period, the inputs (X1-X5) are held low so that all the n-channel transistors 16 are turned OFF, that is, made non-conducting. When thereafter a precharge signal (-PRECHARGE) applied to the gate electrodes of transistors 18 and 20 goes low, p-channel transistor 18 is turned ON and n-channel transistor 20 is turned OFF. This isolates the grounding node N G of the logic group 14 from ground potential but connects the signal node N₀ to the power supply voltage. As a result, the signal node N₀ is charged during the precharge period to the supply voltage. The signal node N₀ is further connected to the input of a CMOS inverter gate 22 consisting of a p-channel transistor 24 and an n-channel transistor 26 serially connected between the power supply voltage and ground potential of the circuits. The inverter input connects node N₀ to the gate electrodes of transistors 24 and 26 and an inverter output Y is taken from the common connection between the two transistors 24 and 26. It should be noted that since at the end of the precharge period, the signal on the signal node N₀ is high, the signal on inverter output Y is low. Since the output signal Y may be further connected as an input signal X i to another logic group the low level on the output Y following precharging of its input logic group is consistent with the requirement that all the input to a subsequent logic group must be at a low level during precharging of this group. The primary inputs Pi to the circuit from register 10 are required to follow this same convention by means of appropriate interface circuits. At the end of the precharge period, the (-PRECHARGE) signal returns to its high level and transistor 20 turns ON and transistor 18 turns OFF. The grounding node N G is therefore reconnected to ground but the signal node N₀ is isolated from the power supply and left floating with the precharge. The primary inputs PI₁ - PI₈ from input register 10 then assume their signal values and are applied during an evaluation phase as input to the various logic modules f with the result that low impedance paths are established through some modules and not through others depending upon the levels of inputs signals Xi selectively applied thereto. If in the present example of a logic module or group shown in Fig.2, a low impedance discharge path from the signal node N₀ to the grounding N G is formed by an appropriate combination of input signals X₁ - X₅, then the precharged voltage node N₀ is discharged through the logic group to a low state. This new low state then switches the CMOS inverter 22 and the output signal Y goes high. If, on the other hand, no discharge path is created, then the output signal Y remains high. The output signal Y may be applied as an input to a logic group of a further precharged stage and, may thus cause a corresponding node of that logic group to discharge and so on through the circuit. Hence, the name domino. As described, a CVS circuit is dynamic in that it required the precharging of the signal node N₀ and the subsequent monitoring of the signal on that signal node N₀, to determine whether it continues to be precharged or is discharged through its associated logic group. As with most dynamic circuits, there is a problem associated with the leakage and fluctuation of the precharge from the signal node N₀. Leakage in a dynamic circuit is a pervasive problem and several solutions have been proposed. For example, see US-A-4,433,257 (Kinoshita), US-A-3,989,955, US-A-4,464,587 (Suzuki) and US-A-4,398,270 (Taylor) One method used to reduce the leakage problem is illustrated in Fig.3. Here a p-channel transistor 28 is connected as a feed-back device between the output and input of the inverter 22 and the power supply. The gate electrode of transistor 28 is connected to the output Y of the inverter 22. Thus, when the signal node N₀ is charged or high, the output signal Y is low, transistor 28 turns ON thus closing the feedback path and solidly connecting the signal node N₀ to the power supply voltage. As a result, any leakage from the signal node N₀ is compensated. However, the feedback arrangement needs to be sufficiently weak such that when the signal node N₀ is positively discharged, transistor 28 does not continue to feed sufficient charge to the inverter 22 to prevent its output signal Y from turning off the transistor 28. A yet more elaborate feedback is illustrated in Fig.4, in which additionally an n-channel feedback transistor 30 is connected as a feed-back device between the output and input of the inverter 22 and ground potential. The gate electrode of transistor 30 is connected to the output Y of the inverter 22. Now, when the signal on the signal node N₀ is discharged or low, the output Y rises, transistor 28 turns OFF, transistor 30 turns ON and solidly connects the signal node N₀ and the input of the inverter 22 to ground potential. Thus charge fluctuations and leakage into the signal node N₀ are compensated for. Similarly to the p-channel feedback arrangement, the n-channel feedback arrangement needs to be sufficiently weak so that its effect can be overcome by a subsequent positive charging of the signal node N₀. Together, the feedback transistors 28 and 30 themselves comprise a weak feedback inverter 32, and are illustrated in block form in Fig.5. The two back-to-back inverters 22 and 32 thus act as a regenerative memory for storing any strong signal impressed upon the signal node N₀. CVS circuits have many advantages but the technology appears to be plagued with spurious signals or glitches. The signal on the signal node N₀ is subject to fluctuations, particularly at the beginning of the evaluation phase. The feedback arrangements described above and illustrated in Figures 3 and 4, are aimed at reducing the effects of these fluctuations. However, in a dynamic circuit, the feedback must be weak for circuit to operate and may not be sufficient to prevent the situation arising where an output signal Y varies momentarily to an incorrect high value. Even if the feedback arrangements subsequently bring the output signal Y to the correct low value, the damage may have already been done in that the transient incorrect value may have already discharged the logic group f of a subsequent stage. Thus, the feedback arrangements alone are inadequate to solve the problem. Summary of the Invention Accordingly, an object of this invention is to provide a cascode voltage switching circuit that has reduced susceptibility to internal noise. A further object of this invention is to reduce the fluctuations without an undue increase in the complexity of the circuit. Viewed from one aspect the invention provides a method of precharging a cascode voltage network having at least one gate group each comprising one or more logic gates connected between a voltage node and grounding node, and having an output node carrying a signal derived from said voltage node, each said logic gate being controlled by an input signal from a primary input or from an output node of another gate, said method being characterised by the steps of holding said input signals at their potential levels prior to precharging, precharging both said voltage and grounding nodes of each of said gate groups to a first potential level, and changing said input signals to turn off all said logic gates at the end of precharging. At the beginning of precharging, a pass transistor between the logic gate and the output inverter is turned OFF, thus isolating the output from the precharging voltage. Both the signal node and the grounding node are then precharged. The continued application of the previously derived output signals to the logic groups during this precharging ensures that all internal capacitive nodes which had been discharged in the previous cycle are fully precharged in the current precharging period. Viewed from another aspect, the invention provides a precharge circuit for performing the method of precharging as described above for a logic block (xi) containing one or more gate devices connected between first and second nodes (N o and N g ) and having input means for cyclically applying combinations of bi-level input signals (P₁-P₈) to control inputs (X₁-X₅) of said gates in order to open or close the gates in accordance with the combination of input signals applied during each cycle, said circuit comprising a first switch connecting said first node to a first value reference potential (supply) operable during pre-charge phase of a cycle to charge the first node to the first reference potential, and further comprising a second switch operable during a subsequent evaluation phase of the same cycle, with a selected combination of bi-level input signals applied to said logic block, to connect the second node to the second reference potential (ground) whereby if, as a result of the applied input signals, a conducting path is established through said logic block, current flows through the logic block with a consequential change in potential of said first node to said second reference potential, and output means for detecting such change characterised by means for maintaining the same combination of input signals applied to said logic block as were applied during the evaluation phase of the immediately preceding cycle during the start of the pre-charge phase of a current cycle, means for changing the input signals as appropriate so that all gates within the logic block are turned off before the end of the current pre-charge phase, and a third switch operable during said pre-charge phase to connect said second node to the first reference potential. Brief Description of the Drawings Figure 1 is a simplified schematic of an exemplary cascode voltage switch circuit. Figure 2 is a schematic illustration of a module in a CVS circuit. Figures 3 and 4 are improvements of the circuit of Figure 2, utilising feedback. Figure 5 is a schematic diagram illustrating the feedback memory of Figure 4. Figure 6 is a schematic illustration of the precharging points of the circuit of Figure 2. Figure 7 is a schematic illustration of one embodiment of the present invention. Figure 8 is a timing diagram for the circuit of Figure 7. Figure 9 and 10 are alternative memory units for the circuit of Figure 7. Detailed Description of The Preferred Embodiments The inventors believe that a major cause of the signal fluctuations in cascode voltage switch (CVS) circuits arises from redistribution of the precharge dependent upon the signal inputs of the prior evaluation phase. This effect will be explained with reference to Figure 6, which illustrates some details omitted from Figure 2. The precharging of the signal node N₀ implies that there is a substantial capacitance 34 associated with that node. In MOS technology, there is often enough parasitic capacitance associated with the long interconnections between the signal node N₀, the upper precharge transistor 18, and the inverter 22 so that the capacitance 34 does not need to be separately provided. In the precharging phase, the capacitance 34 is positively charged and is discharged in the evaluation phase if a conducting path is formed through the logic group 14 to ground. However, the capacitance 34 is not the only parasitic capacitance in the circuits. There is additional capacitance 36, 38 and 40 at the points between the logic gates 16. However, according to the design convention for CVS circuits, all the inputs X₁-X₅ are low during the precharging period so that the associated transistors 16₁-16₅ are turned off during precharging. As a result, the parasitic capacitances 36-40 within the logic group 14 are not accessible and are not precharged. For the most part, the amount of charge on the capacitances 36-40 within a logic group at the end of the precharging period is determined by the amount to which they were charged at the end of the prior evaluation period. This amount of charge, in turn, is determined by the values of the input signals X₁-X₅ to the current logic group which were applied in the prior evaluation period. For example, assuming that the internal capacitance 36-40 were fully charged before the prior evaluation period, then if all the input signals X₁-X₅ were low (binary '0') in the prior evaluation period, the internal capacitance 36-38 remain fully charged for the current evaluation period. If, on the other hand, all the input signal X₁-X₅ in the prior evaluation period were high (binary 1), then all the internal capacitances 36-40 of the group will have been discharded. Furthermore, different combinations of input signals X₁-X₅ in the prior evaluation phase can cause different combinations of the internal capacitances 36-40 to be discharged. The importance of precharging inherent capacitance has been discussed in US-A-3,778,782 (Kitagawa) and in US-A-4,200,917 (Moench). In order to understand the effect of the varying value of the internal capacitances the different situations will be considered in some detail. In the subsequent evaluation phase of a logic group following precharging, the precharge on the signal node N₀, that is, the precharge on the capacitance 34, may be discharged to ground if the input signals X₁-X₅ turn on sufficient transistors 16 to form a conducting path to ground. In this case, the precharge on the signal node N₀ is discharged to ground through the conducting path of transistors, although the transition time may depend upon the amount of charge on the internal capacitances 36-40. If the input signals X₁ and X₂ are low during the evaluation phase, then all the precharge remains on the signal node N₀ and a solid high signal is presented to the inverter 22. The real problem arises when the combination of inputs X₁-X₅ are such as not to form a conducting path to discharge the node N₀ but nonetheless one or more of the transistors adjacent the node N₀, such as transistors 16₁ or 16₂ are ON. As a result, the capacitance 36 or 40 or possibly both are put in parallel with the capacitance 34. If the internal capacitance 36 or 40 remains charged following the previous evaluation phase, then the problem should be relatively small because all the capacitances have been charged by similar voltages. However, if the internal capacitance 36 or 40 has been discharged in the prior evaluation phase, then the precharge on the capacitance 34 is redistributed to the discharged capacitance 36 or 40 or possibly both. This redistribution of the precharge causes the voltage on the signal node N₀ to dip. The amount of this dip is thus dependent upon the input signals X₁-X₅ of the prior evaluation phase as well as the input signals X₁-X₅ of the current evaluation phase, and thus is difficult to predict or control. The feedback around the inverter 22, as shown in either Figures 3 or 4, can compensate for this dip and bring the signal on the signal node N₀ back up to a solid high as long as the dip is not too severe. However, as previously mentioned, the feedback must be weak so that a transient dip appears inevitable in view of the dynamic nature of a CVS circuit. It is possible that the feedback eventually brings the high signal on the signal node N₀ back to its correct value, but, in the meantime, the output signal Y may have changed enough to have caused a logic group f of the following stage to have discharged. Once this following logic group has been discharged, the application of the correct input signals to that stage, intended to keep that logic group charged, cannot recharge the signal node N₀ of the following group. As a result, transient signal errors become permanent by the domino nature of a CVS circuit. The invention uses this understanding of the problem to reduce fluctuations by the following method. The output signal Y is memorised and then held during a part of the precharging period at the level it had in the prior evaluation phase. That is before the current precharging period. Then whilst the value is held, both the signal node N₀ and the ground node N G of the logic group are both precharged. Since precharging of all the logic units is conducted simultaneously, the inputs Xi to all the logic groups are held at the values of the previous evaluation phase. However, as will be seen the inputs Xi are forced to the low level before the end of the precharge phase as is required. The implementation for the CVS module of Figure 4 is shown in Figure 7. A (-PRECHARGE) signal, having a slightly different function than the (-PRECHARGE) of Figures 2 and 6, directly controls the gate of an n-channel pass transistor 42 connecting the signal node N₀ to the input to the inverter 22. Thus, at the beginning of the precharge period, with (-PRECHARGE) signal low, transistor 42 is turned OFF and the signal node N₀ is isolated from the inverter 22 and the weak feedback inverter 32. As previously mentioned, the combination of the inverters 22 and 32 act as a memory device. With the memory device thus isolated from the precharging, the value of the output signal Y of the prior evaluation phase is held during the current precharging period. This condition applies to the output of all CVS modules. The (-PRECHARGE) signal is passed, as shown in Figure 8, through two inverters 44 and 46, to form a delayed precharged signal on line C₂ which is applied to the upper and lower precharge transistors 18 and 20 in the usual way and additionally to a further precharge p-channel transistor 48 connected between the power supply and the grounding node N G of the logic group 14. The delayed low level of the (-PRECHARGE) signal turns transistor 20 OFF and transistors 18 and 48 ON. As a result, following the downward transition of the delayed (-PRECHARGE) signal on line C₂, the signal node N₀ having been previously isolated from the inverters 22 and 32 to which it is normally connected is then connected to the power supply via the transistor 18 for precharging. At the same time, the grounding node N G is isolated from ground by the transistor 20 turning OFF and itself connected to the power supply for precharging via the transistor 48. Thus, both nodes N₀ and N G of the logic group 14 are precharged to the supply voltage. During this delayed precharging period, the input signals X₁-X N are held at their respective values of the prior evaluation phase by memory units of other logic modules corresponding to the inverters 22 and 32. Primary inputs to the CVS circuits must follow the same convention which could be accomplished by attaching a pass gate 42 and inverters 22 and 32 to the primary inputs PI i . In summary, certain combinations of input signals Xi of the prior evaluation phase caused certain of the transistor 16 to turn ON and thus discharge their associated internal capacitances 36-40. The discharging could either have been directly downward to the grounding node N G or could have first followed an upward path to the signal node N₀ before reversing to a fully conductive path to the grounding node N G . In some types of CVS circuits, it may be possible that more complex discharging paths are formed. Thus, whatever discharging paths to the nodes N₀ and N G were formed during the prior evaluation phase are also maintained at least for a period in the current precharging. However, both of the nodes N₀ and N G are held at the precharging voltage during precharging so that whatever internal capacitances 36-40 were discharged during the prior evaluation are once again precharged. The precharging of all capacitances is to the same supply power voltage. The relative timing of the (-PRECHARGE) signal and the delayed (-PRECHARGE) signal on line C₂ is shown in Figure 8. The important point is that the delay is substantially less than the duration of either pulse. A value of 600 ps for Δt would be reasonable although its best value depends on the rest of the circuit. When the (-PRECHARGE) signal goes high, the first occurrence is for the pass transistor 42, to be turned ON. Immediately, the precharge now existing on the signal node N₀ forces the output signal Y to a low level. As a result, all the input signals Xi go to a low level because the same convention is applied to these inputs. Thus all the transistors 16 in all logic groups such as the logic group 14 are turned OFF for the last part of the precharging phase by the delayed signal precharge signal still on line C₂. This prevents a memorised conducting path through a logic group from erroneously discharging the precharged signal node N₀ if the various transitions are not timed exactly. Finally, after delay Δt the delayed (-PRECHARGE) signal on line C₂ also goes high to discontinue the precharging process and turn OFF transistors T18 and T48 and turn ON transistor 20 thereby reconnecting the grounding node N G to the ground potential ready for the subsequent evaluation phase. Then, following this precharging phase, the application of the current set of the primary inputs PI i will initiate the domino effect upon fully precharged logic groups 14. The internal capacitances 36-40 have been precharged to the same voltage as the capacitance 34. As a result, there is no redistribution of precharge. The circuit of Figure 7 shows that the delayed precharge signal on line C₂ is generated locally, that is, in the neighbourhood of the logic group 14, directly from the (-PRECHARGE) signal. This approach necessitates two additional inverters 44 and 46 for each logic group 14. It is possible for both the precharge signal (-PRECHARGE) and the delayed version on line C₂ to be generated at one point on an integrated circuit and then both to be distributed to all of the logic groups 14 on the integrated circuit. This central generation of both precharging signals of course necessitates additional interconnections. However, the routing for both sets of interconnections can be the same so that the complexity is not greatly increased. Furthermore, if the two interconnections are routed in parallel, any time skew affecting one of the precharging signal affects the other. As a result, the amount of delay Δt is easily maintained over the long interconnections. The embodiment shown in Figure 7 used two back-to-back inverters 22 and 32 as a fully regenerative memory unit. However, it should be remembered that there is some associated parasitic capacitance 50, as shown in Figure 9, on the interconnection 52 between the pass transistor 42 and the inverter 22. Furthermore, the main leakage associated with this interconnection 52 is to ground. That is, the capacitance 50, for the most part, adequately holds a low signal but is somewhat undependable for a high or charged signal. Accordingly, the feedback gate 30 connected between the interconnection 52 and ground in the prior art circuit can be dispensed with. The required memory function can be adequately fulfilled by the p-channel feedback transistor 28, the capacitance 50 and the inverter 22. It is also possible, as shown in Figure 10, to completely dispense with both feedback transistors in the memory unit since the capacitor 50 itself is an adequate memory unit. Admittedly, it is somewhat leaky, particularly for high level signals but as long as the capacitance 50 is required to memorise the signal only for a time shorter than its leakage time, it alone is satisfactory. The leakage time can be lengthened by intentionally including a large capacitance 50 that has relatively low leakage compared to that of the interconnection 52. Nonetheless, the capacitor 50 and the interconnection 52 do leak so that the circuit of Figure 10 is dynamic. As a result, there will be a minimum operating frequency between the precharging periods to ensure that the memory unit of Figure 10 does not decay to an incorrect value. In the embodiments of Figures 9 and 10, it is of course assumed that the rest of the circuit of Figure 7 is maintained.

Claims: A method of precharging a cascode voltage network having at least one gate group each comprising one or more logic gates connected between a voltage node and grounding node, and having an output node carrying a signal derived from said voltage node, each said logic gate being controlled by an input signal from a primary input or from an output node of another gate, said method being characterised by the steps of: holding said input signals at their potential levels prior to precharging, precharging both said voltage and grounding nodes of each of said gate groups to a first potential level, and changing said input signals to turn off all said logic gates at the end of precharging. A precharge circuit for performing the method of precharging as claimed in claim 1 for a logic block (xi) containing one or more gate devices (161-165) connected between first and second nodes (N o and N g ) and having input means for cyclically applying combinations of bi-level input signals (P₁-P₈) to control inputs (X₁-X₅) of said gates in order to open or close the gates in accordance with the combination of input signals applied during each cycle, said circuit comprising a first switch (18) connecting said first node to a first value reference potential (supply) operable during pre-charge phase of a cycle to charge the first node to the first reference potential, and further comprising a second (20) switch operable during a subsequent evaluation phase of the same cycle, with a selected combination of bi-level input signals applied to said logic block, to connect the second node to the second reference potential (ground) whereby if, as a result of the applied input signals, a conducting path is established through said logic block, current flows through the logic block with a consequential change in potential of said first node to said second reference potential, and output means (22) for detecting such change characterised by means for maintaining the same combination of input signals applied to said logic block as were applied during the evaluation phase of the immediately preceding cycle during the start of the pre-charge phase of a current cycle, means for changing the input signals as appropriate so that all gates within the logic block are turned off before the end of the current pre-charge phase, and a third switch (48) operable during said pre-charge phase to connect said second node to the first reference potential. A precharge circuit as claimed in claim 2, in which said output means includes an output circuit (22) operable in response to potential changes of said first node to generate bi-level output signals, one level output signal indicating that the first node is at the first potential and the other level indicating that the first node is at the second potential, said output means further comprising a fourth switch (42) connected between the first node and the output circuit and operable to disconnect the output circuit from the first node for a predetermined period, said period commencing immediately prior to the start of a precharge phase of the associated logic block but being re-connected before the end of the precharge phase, said output circuit including memory means sufficient to maintain the integrity of the current output signal level of the output circuit during the period of isolation from said first node. A logic network comprising a plurality of logic blocks associated output circuits as claimed in claim 3, the input signals to the logic blocks being supplied from the outputs of a loadable input register (10) or from the output of a preceding logic block circuit, the output signal from the output circuits being supplied to the input stages of an output register (12) or to control inputs of a succeeding logic block circuit. A precharge circuit as claimed in claim 3, in which said first and third switches are MOS transistors of one conductivity type and said second and fourth switches are MOS transistors of the opposite conductivity type, and including control means to supply a bi-level control signal to the gates of said transistors whereby when the control signal is at one level the MOS transistors of one conductivity type are switched ON and when the control signal is at the other level, the MOS transistors of the opposite conductivity type are switched ON, the arrangement being such that the control signal applied to the gate of said fourth transistor is shifted in time with respect to the corresponding control signal applied to said first second and third transistors such that the signal level causing said fourth transistor to switch off occurs in advance of the corresponding signal level causing the second transistor to switch off and the first and third transistor to switch on, and consequently the signal level causing the fourth transistor to switch ON occurs in advance of the corresponding signal level causing the second transistor to switch ON and the first and third transistors to switch OFF. A precharge circuit as claimed in claim 5, in which the first reference potential is the supply potential, the second reference potential is ground potential, the first and third transistors are p-channel devices and the second and fourth transistors are n-channel devices, and in which the bi-level control signal is applied directly to the gate electrode of the fourth transistor and via a delay (44, 46), small with respect to the pulse length of the bi-level signal, the gate electrode of the first, second and third transistors. A precharge circuit as claimed in claim 6, in which the gates comprising a logic block consists of n-channel MOS transistors and the output circuit operates to produce at its output a signal level the inverse of the signal level on said first node whereby a high level signal on the output indicates that the applied combination of input signals to said logic block during an evaluation phase has matched the internal logic representation of said block. A precharge circuit as claimed in claim 7, in which said output circuit incorporating memory comprises a first buffer amplifier (22) having an input connected to the fourth transistor and a second buffer amplifier (32) being weaker than said first buffer amplifier and having its output connected to the output of said fourth transistor and having its input connected to the output of said first buffer amplifier which is also the output (Y) of the circuit. A precharge circuit as claimed in claim 8, in which said second buffer amplifier consists of an MOS transistor (28 or 30) connected between the output of said fourth transistor and one or other of said first or second reference potentials and its gate electrode connected directly to the output of said first buffer amplifier.

Inventors: LING, DANIEL TAJEN, OKLOBDZIJA, VOJIN G, RAVER, NORMAN
Applicants: INTERNATIONAL BUSINESS MACHINES CORPORATION
Application year: 1986
Main classification: H03K19
IPC codes: H03K17, H03K19