COMBINED CROSS POLARIZATION INTERFERENCE CANCELLATION AND INTERSYMBOL INTERFERENCE EQUALIZATION FOR TERRESTRIAL DIGITAL RADIO SYSTEMS

Abstract: Data-aided equalization and cancellation in digital data transmission over dually polarized fading radio channels is disclosed. The present invention uses decision feedback structures (150) with finite-tap transversal filters (100). Sub­ ject to the assumption that some past and/or future data sym­ bols are correctly detected, formulas and algorithms for eval­ uating the least mean-square error for different structures are disclosed. In a sequence of curves, the performance of various structures for a particular propagation model and several fad­ ing events are evaluated and compared.

Description: This invention relates to an apparatus and a method for a combined cross polarization cancellation and intersymbol interference equalization according to the introductory portions of claims 1 and 4 respectively. Transmission of M-state quadrature amplitude modulated (QAM) signals via orthogonally polarized carriers results in the conveyance of increased information within a given bandwidth, resulting further in economic advantages. The main obstacle in the way of realizing these advantages is the unavoidable presence of cross polarization interference (CPI) between the dually polarized signals which arise due to factors such as multipath fading, antenna misalignments and imperfect waveguide feeds. The theory of linear and decision feedback equalization/cancellation, to mitigate the effects of intersymbol interference (ISI) and noise in the transmission of a single digital signal, is well established in an article entitled "Digital Communication over Fading Radio Channels", Bell System Technical Journal, Vol. 62, No. 2, Part 1 (1983), pages 429-459 by G. J. Foschini and J. Salz. But a combination of data aided compensation of CPI cancellation and ISI equalization in the presence of noise in the aforesaid manner has not been solved heretofore. From a paper by J. Namiki et al. "Adaptive Receiver for Cross-Polarized Digital Transmissions", International Conference on Communications, Conference Record, Denver, Colorado, June 14-18, 1981, Vol. 3, pages 46.3.1-46.3.5, there is prior known a receiver having a decision feedback equalizer and a cross polarization canceller which both are using transversal filters. Starting from that prior art the invention is characterized in claims 1 and 4 respectively. Summary of the Invention In accordance with the illustrative embodiment of the present invention, the problem of cross polarization interference (CPI) cancellation and intersymbol interference (ISI) equalization is solved by using data aided decision feedback techniques. Most importantly, finite tap transversal structures which are implemented adaptively are used in a receiver. A window sample of the signal that is received is processed in two sections. The first section is equalized and the interference in the remaining section is canceled. Furthermore, the taps are not co-located. That is the equalizer taps operate over a range where the canceler taps are not operative. Likewise, the canceler taps operate over the remaining range where the equalizer is not operative. The receiver configuration is based on a matrix structure from the theory of optimal detection. The structure is optimal when the mean square error (MSE) of the received signal from the ideal signal is minimized. The optimal structure comprises a matrix matched filter in cascade with a transversal filter and an intersymbol interference and cross polarization interference canceler. The canceler uses detected data symbols to estimate the interference to be canceled. Data aided operations presume correct knowledge of detected data symbols. A dually polarized channel is modeled by a particular four by four real matrix impulse response or its Fourier transform followed by additive noise. The two by two block diagonal elements of this matrix represent the co-polarized (in line) responses, while the off-diagonal two by two block entries represent cross coupled and cross polarized interfering responses. Each matrix channel characterizes a snapshot of a multipath fading event which in the presence of noise limits the achievable error rate of the receiver for a given data rate. For a reasonable co-polarized and cross polarized propagation model and a severe centered fade, 40 dB notch depth with a secondary signal delay of 6.3 ns, over an approximately 22 MHz channel bandwidth, the performance of transversal filters with a finite number of taps deployed in a decision feedback canceler structure is substantially (6 dB) better than linear equalization, and the difference can be up to 10 dB for offset fades. It can be shown that a 3 dB increase in MSE translates into about 1 bit/sec/Hz decrease in data rate efficiently at a fixed error rate or an order of magnitude increase in outage probability. Hence, linear equalization may not be adequate in deep fades. Decision feedback/canceler structures achieve the ultimate matched filter bound with only nine matrix taps provided that error propagation is neglected. Nine linear equalizer taps essentially achieve the performance of the infinite-tap linear equalizer. This method is, of course, free of error propagation. For milder centered fades, 20 dB depth with a secondary signal delay of 6.3 ns the linear equalizer configuration with nine taps is only 1 dB inferior to the decision feedback structure over a 22 MHz channel. If the channel bandwidth is increased to 40 MHz, however, the performance of the linear equalizer is worse than that of the decision feedback structure by 2.2 dB and the difference can be up to 3 dB for offset fades. Decision feedback/canceler configurations are less sensitive to timing phase than linear structures. Brief Description of the Drawing FIG. 1 shows a block diagram of a dual polarized system; FIGS. 2, 3 and 4 show details of a receiver embodying the present invention; and FIGS. 5 through 11 show curves for evaluating the performance of the aforesaid receiver embodying the present invention. Detailed Description Referring to FIG. 1, there is shown a prior art transmission system which utilizes a dually polarized digital radio communication channel supporting two independent quadrature amplitude modulated (QAM) data signals. The system shows an ideal QAM modulator 10, and demodulator 50. Four independent synchronous data signals S 1v (t),S 1h (t), where 1 = 1, 2 and which are represented generically as S 1v (t) = Σ / n a 1vn g(t-nT), where 1 = 1, 2 and amplitude modulate two linearly polarized carrier waves in quadrature at multipliers 12, 14, 16, and 18. The vertically modulated signal, is transmitted over the vertically polarized channel 21, while the horizontally modulated signal, is transmitted over the horizontally polarized channel 23. The carrier frequency is ω₀ and the real data symbols a 1vn , 1 = 1,2 and a 1hn , 1 = 1, 2 , where -∞ < n < ∞ are assumed to be independently drawn from a lattice of points with odd integer coordinates. The QAM constellations associated with equations (2) and (3) are, therefore, rectangular. The scalar pulse, g(t), is selected to satisfy limitations on transmitted power and bandwidth. The individual transmission channels are characterized by bandpass impulse responses or by their Fourier transforms, To accommodate coupling between the polarized channels 21 and 23, two pairs of impulse responses, one pair 30, 32 associated with the co-channel and the other pair 34, 36 associated with the cross-channels, are used to characterize completely the medium. Two independent noises, γ iv cos(ω₀t) + γ qv sin(ω₀t) and γ ih cos(ω₀t) + γ qh sin(ω₀t) are added to the cross polarized signals, respectively, at multipliers 62 and 64. These multipliers are located at receiver 50 of FIG. 1 where the signal plus noise is then coherently demodulated. It is convenient to view the linear system as a four input port, four output port network and characterize it as a four by four matrix impulse response or its Fourier transform which is the overall system frequency response. The cross polarized signals on lead 63 having been mixed with the aforesaid noise signals are bifurcated into paths 51 and 53 and then delivered, respectively, to mixers 52 and 54 where the signals are mixed with the delayed carriers cos(ω₀t + Θ) and sin(ω₀t + Θ). Likewise, the cross polarized signals on lead 65 having been mixed with the aforesaid noise signals are bifurcated into paths 55 and 57 and then delivered, respectively, to mixers 56 and 58 where the signals are mixed with the carrier signals cos(ω₀t + Θ) and sin(ω₀t + Θ). These signals are then low pass filtered through elements 72, 74 and 76, 78 and then sampled by the sets of samplers 82 and 84; and, 86 and 88, respectively. Signals on output leads 81, 83 correspond with input leads 11, 13 and output leads 87, 89 correspond with input leads 17, 19. Referring to FIG. 2, there is shown a block diagram useful in disclosing the present invention and comprising a transmitter and channel 10 and receiver 111. Receiver 111 comprises intersymbol interference equalizer 100, a linear matrix filter, the output from which is supplied to detector 140. Detector 140 compares the signal generated internally input thereto with an expected signal and produces an error from which a mean square error signal is generated and this error signal is transferred to cross polarization interference canceler 150 as well as to intersymbol interference equalizer 100. The characteristics of the aforesaid linear matrix filter 100 are selected so as to minimize the total mean square error. Canceler 150 produces adjusting signals proportional to the interference caused by cross polarization and delivers these signals to subtracters 102 and 104. After the signals which adjust for cross polarization interference are subtracted from the filtered estimates, the resulting signals are substantially free from interference. Further detail of receiver 111 is shown in FIGS. 3 and 4. Referring to FIG. 3, there is shown an input signal I₁ on rail 71 and Input signal Q₁ on rail 73. These two signals correspond with the signal on lead 63 of FIG. 2 prior to bifurcation. In order to effect comparison with FIG. 1, identical numbers are used where convenient. Likewise, the input signals I₂ on rail 75 and Q₂ on rail 77 correspond with the signals on lead 65 of FIG. 2 prior to bifurcation. Input signals are sampled by samplers 82, 84, 86 and 88 prior to delivery to the feedforward transversal filters 110, 112, 114 ... 139 for a digital system. For an analog system, the samplers appear after the filters as shown in FIG. 1 hereinabove. Input signal I₁ is delivered to feedforward transversal filters 110, 114, 126 and to 128. Likewise, each of the other input signals is delivered to four feedforward transversal filters. The taps of each feedforward transversal filter are adjusted adaptively by error signals from a detector to be described hereinbelow. The output from feedforward transversal filter 110 is fed to mixer 105. Because the other signals are coupled with this input signal I₁ during transmission, the other input signals, Q₁, I₂, and, Q₂, must be decoupled. Decoupling of these signals from input signal I₁ is effected by feeding the output signals from feedforward transversal filters 112, 118 and 122 to mixer 105. Likewise, each of the signals is decoupled from the others. In the preferred embodiment sixteen transversal filters are used. This is apparent because there are four input signals. Details of design are not disclosed in this application because such design is trivial to one skilled in this art. The decoupled signals must be adjusted further by the output signals from the feedback transversal filters to be disclosed with reference to FIG. 4 hereinbelow. This is effected in mixers 106, 108, 96 and 98. The output from mixer 106, for example, is then sent to detector 141. The input to detector 141 is compared with the output therefrom in comparator 142 and the output is sent as error signal ES₁ to a plurality of filters for adaptive adjustment of the taps therein. Error signal ES₁ is sent, for example, to feedforward transversal filters 110, 114, 126 and 128 and to feedback transversal filters 151, 152, 155 and 156. Likewise, the taps of the remaining filters are adjusted adaptively by error signals from comparators 144, 146, and 148. The output signals from the detectors are substantially free from CPI and ISI. Referring to FIG. 4, there is shown canceler 150 of FIG. 2 comprising a plurality of feedback transversal filters 151, 152, 153 ... 166 each being adjusted by an error signal from detector 140 of FIG. 2. Each feedback transversal filter has input thereto an output signal from detector 140. Thus, filter 151, for example, has input thereto the output signal I₁ from detector 141 of FIG. 3. The output signal from feedback transversal filter 151 is delivered to mixer 167. Also delivered to mixer 167 are output signals from filters 153, 159 and 161 so that these signals may be decoupled. The output from mixer (167) is then delivered to mixer 106 of FIG. 3. Likewise, the other feedback filters of FIG. 4 cause the remaining signals to be decoupled. Referring to FIG. 5, there is depicted the normalized least mean square error (MSE₀ ) as a function of the number of canceler taps, Q, when a 40 dB centered fade over a 22 MHz channel band is applied to both polarized signals. The linear equalizer in this case possesses an infinite number of taps. The case of pure linear equalization, N₁ = N₂ = 0, that is, no cancellation, exhibits the largest MSE₀ degradation relative to the asymptotic matched filter bound. This is due to the noise enhancement experienced by the linear equalizer during deep fades. When both causal and anticausal canceler taps are present, all the curves rapidly approach the matched filter bound for a finite constant coupling, K i -20dB, i = 1,2,4,5. The curve for a decision feedback type canceler starts at ideal decision feedback equalizer normalized MSE₀ and approaches the asymptotic value with two anticausal taps. The finite window size canceler curve starts at the linear equalizer case, N₁ = N₂ = 0, and reaches the matched filter bound asymptotic value with a total of four causal/anticausal taps. Finally, when no anticausal taps are employed, the curve asymptotically approaches the ideal decision feedback case with only two causal taps. Referring to FIG. 6, there is depicted similar results as in FIG. 5 for the case when the centered fade notch depth is reduced to 20 dB over a 22 MHz channel. As can be observed the linear equalizer, N₁ = N₂ = 0, performance is improved. In both FIGS. 5 and 6, the fade notch is located at the band center. In both cases, because the receive filter matches the overall channel and transmitter, an offset fade notch does not have a serious impact on the results for the same fade notch depth. Referring to FIGS. 7 and 8, there is depicted in each, the achievable MSE₀ when the linear equalizer has a finite number of taps. The fade notch in FIG. 7 is centered but in FIG. 8 is offset from the band center. For ease of presenting the results, fade notch offset from the band center is expressed in terms of the ratio of the fade notch distance from the band center to the channel equivalent baseband bandwidth in percentage. In FIG. 8, the fade notch is offset by 69 percent over a 22 MHz channel, that is, an offset of 7.6 MHz from the band center. As observed from FIG. 7, a total of nine taps, including the center tap, are required to achieve the asymptotic matched filter bound when decision feedback taps are present. The same asymptotic performance can be achieved no matter how the nine synchronously spaced taps are deployed between the linear equalizer and the canceler as long as the canceler operates in a decision feedback mode. This is because the equalizer and canceler tap windows complement one another; therefore, since the taps do not overlap, for the same number of taps, the performance remains almost the same in the decision feedback cases. An important configuration is seen when the linear equalizer operates only on the main lobe of CPI by means of its center matrix taps, M₁ = M₂ = 0. It is clear that as long as the canceler window is sufficiently wide, a main lobe CPI canceler can achieve the asymptotic matched filter bound. The curves indicate that deep fades degrade the linear equalizer, N₁ = N₂ = 0, performance significantly. Previous studies showed that every 3 dB degradation in MSE₀ translates into a loss of 1 bit/sec/Hz of data rate efficiency. Hence, linear equalization may not provide adequate rate efficiency in deep fades. In FIG. 8, similar curves are depicted, as in FIG. 7, but for a 40 dB fade with the notch frequency offset by 69 percent. Improved performance turns out to be due to the particular notch position as will be brought out in the discussion of FIG. 10. FIG. 9 illustrates a similar set of curves for a 20 dB centered fading of dually polarized signals over both a 22 MHz and a 40 MHz channel. As can be observed, the linear equalizer, N₁ = N₂ = 0, performance improves because of the decreased fade depth. Over the wider channel band, however, the degradation over decision feedback is more, as expected. This is due to the wider channel band over which the same fade notch causes more dispersion. The degradation amounts to 2.2 dB loss of MSE₀ comparing to matched filter bound, that is, roughly 1 bit/sec/Hz loss of data rate efficiency, and the loss can even be more for offset fades as will be seen in FIG. 10. Hence, even with more typical fades the use of the linear equalizer can be troublesome over a 40 MHz channel. Finally, to compare some of the techniques described earlier in terms of their sensitivity to fade notch offset, the normalized MSE₀ is plotted In FIG. 11 as a function of fade notch position, which, as explained hereinbefore, is expressed here in terms of the ratio of the fade notch distance from the band center to the channel equivalent baseband bandwidth. Consider the following structures: a. A linear equalizer with M₁ = M₂ = 4 N₁ = N₂ = 0 (no cancellation) b. Center tap only linear equalizer/finite window canceler with M₁ = M₂ = 0 N₁ = N₂ = 4. The sensitivity of the linear equalizer to offset fades is quite pronounced. The center tap equalizer with a finite window canceler exhibits a very small sensitivity to offset fades. The degradation of MSE₀ for some offset fades can be explained considering the fact that these fades cause cross polarization of the imaginary part of a complex QAM signal into its real part, and for a particular notch offset frequency within the band, the coupling reaches its maximum. Therefore, the MSE₀ versus fade notch offset curves exhibit this phenomenon. In dually polarized systems, as in the case of the problem at hand, this is even more pronounced than in single signal transmission, because in the 4 x 4 system under offset fading there is coupling of three interfering data streams into the fourth one. A decision feedback type canceler structure, by canceling the major contributions to CPI and ISI and with a lesser noise enchancement, exhibits an improved performance compared to the linear equalizer. Note that the curves in FIG. 11 have all been obtained under optimum timing conditions. Referring to FIG. 11, there is shown a normalized MSE₀ superimposed by the normalized MS-EC of the received signal curve before equalization or cancellation as a function of sample timing offset from the optimum timing used to investigate the sensitivity of the two structures to timing phase. This is done for a severe fade, namely, a 40 dB fade with a notch frequency offset by 34.5 % over a 22 MHz channel. It is clear that the finite linear equalizer is much more sensitive to timing phase than the decision feedback type. The optimum timing reference is established based on minimizing the MS-EC of the received signal in presence of fading, before CPI and ISI cancellation. Hence, after cancellation occurs, this timing reference may not be the one that minimizes the canceler output MSE₀, and indeed the linear equalizer curve of FIG. 11 indicates this fact. The MS-EC curve has a minimum at the optimum timing reference. The sensitivity of the matrix linear equalizer to timing phase can be reduced by applying half-a-baud spaced taps, that is, by deploying fractionally spaced taps. It is to be noted that a decision feedback timing method may prove more robust. But such a method is more complex. The degradation seen in FIG. 11 of the MSE₀ is partly attributed to the asymmetric amplitude and delay responses of the fading channel which in the presence of a nonzero roll-off of the shaping filters cause destructive addition of aliases. So, reducing the roll-off of the shaping filters can also improve this situation.

Claims: Apparatus for combined cross polarization interference cancellation and intersymbol interference equalization of signals (I₁ , Q₁; I₂ , Q₂) received over dispersive terrestrial digital radio transmission channels said signals being in dual polarization mode, comprising means (82, 84, 86, 88) for obtaining samples of said signals having cross polarization interference and intersymbol interference, equalizer means (100) for equalizing said samples and for supplying as outputs equalized samples including a first set of transversal filters (110, 112, 114, ... 138, 139) being adaptively adjusted by first error signals (ES₁ to ES₄), canceler means (150) for canceling said cross polarization interference including a second set of transversal filters (151 to 166) being supplied with output signals (I and₁, Q and₁; I and₂, Q and₂), and being adaptively adjusted by said first error signals (ES₁ to ES₄) for generating second error signals (E, F, G, H), combining means (106, 108; 96, 98) for algebraically combining said equalized samples and said second error signals (E, F. G, H) to generate cross polarization canceled versions of said equalized samples, detector means (141, 143, 145, 147) for detecting said versions of said equalized samples and for deriving said output signals (I and₁, Q and₁; I and₂, Q and₂ ), and comparator means (142, 144; 146, 148) for deriving said first error signals (ES₁ to ES₄ ) by comparing said versions of said equalized samples to said output signals, characterized by said first set of transversal filters (110, 112, 114, ... 138, 139) having taps at selected first intervals and said second set of transversal filters (151 to 166) having taps at selected second intervals, said first intervals being different from said second intervals. The apparatus as defined in claim 1 characterized in that said first set of transversal filters includes a plurality of feedforward transversal filters (110, 112, 114, ... 138, 139) and said second set of transversal filters includes a plurality of feedback transversal filters (151 to 166). The apparatus as defined in claim 1 characterized in that said comparator means (142, 144 146, 148) derives said first error signals (ES₁ to ES₄ ) by obtaining a means square error signal between said output signals and said versions of said equalized samples. A method for combined cross polarization interference cancellation and intersymbol interference equalization in dually polarized signals received over a plurality of dispersive terrestrial digital radio transmission channels comprising the steps: a. obtaining estimates of said signals in each of said channels, b. equalizing said estimates by employing a first set of transversal filters having a plurality of delay taps and being adaptively adjusted by first error signals to generate equalized estimates, c. generating second error signals by employing a second set of transversal filters having a plurality of delay taps, being adaptively adjusted by said first error signals and being supplied with output signals, d. canceling said interference by algebraically combining said second error signals and said equalized estimates, to generate canceled versions of said equalized estimates, e. detecting said versions of said equalized estimates to generate said output signals, and characterized by f. comparing said versions of said equalized estimates and said output signals to generate said first error signals, said delay taps of said first set of transversal filters and said delay taps of said second set of transversal filters being selected at delay intervals which do not overlap one another. The method as defined in claim 4 characterized in that said first set of transversal filters includes a plurality of feedforward transversal filters and said second set of transversal filters includes a plurality of feedback transversal filters.

Inventors: KAVEHRAD, MOHSEN, SALZ, JACK
Applicants: AMERICAN TELEPHONE AND TELEGRAPH COMPANY
Application year: 1986
Main classification: H04B7
IPC codes: H04B7, H04B3, H04L25, H04J11