Pulse-width modulation drive circuit

Abstract: A pulse-width modulation drive circuit for a load such as a motor (M) such as used in a compact disc player or the like, in which losses in diodes (D₁,D₂) used to absorb reverse EMF are substantially eliminated. For this purpose, the pulse widths of pulse signals used to on-off control drive current flow through the load (M) are controlled so as to eliminate the diode losses. The circuits (15, 16) of the various embodiments of the inven­ tion are designed so as to be easily implemented in integrated circuit form.

Description: The present invention relates to a switching drive circuit, and, more particularly, to a circuit which drives a load by switching it on and off in response to a predetermined pulse signal. Motors and other loads can be driven by several methods, among which the two-way switching drive method is quite common. This method has the advantages of low loss and reduced power consumption, and is particularly effective for driving loads such as motors in battery-powered devices, which may be portable or can be installed on vehicles. A configuration of a conventional two-way switching drive circuit is shown in Fig. 1A. The circuit includes a pnp transistor Q₂₀ and an npn transistor Q₂₁, which are connected to each other at their collectors, and a pnp transistor Q₂₂ and an npn transistor Q₂₃, which are also connected to each other at their collectors. A motor M is connected between the two common collector junctions. When a negative pulse signal is applied to the base of transistor Q₂₀ while a positive pulse signal is being applied to the bae of Q₂₃, a drive current will flow in the motor M in the direction indicated by a solid-line arrow, causing the motor to rotate in the forward direction. When, on the other hand, a negative pulse signal is applied to the base of transistor Q₂₂ and a positive pulse signal to the base of transistor Q₂₁, a drive current will flow in the motor M in the direction indicated by a dashed-line arrow, causing the motor to rotate in the reverse direction. As shown in Fig. 1A, one terminal of the motor M is connected to a power supply V cc via a diode D₁₀, while the other terminal of the motor is connected to V cc via another diode D₁₁. Both diodes serve to absorb reverse electromotive force. In terms of an equivalent circuit, the motor M consists of two components, a resistance and an inductance. Part of the energy of the drive current flowing in the motor M is consumed in the resistance as driving energy, and the remainder is stored in the inductance. The stored energy produces a reverse electromotive force when the applied drive pulse is removed, and, after flowing through a closed loop including the diode D₁₀ or D₁₁, the energy will be consumed as driving energy in the resistance. Therefore, theoretically, all of the energy that is generated in the circuit will be consumed as driving energy and the circuit has a linear input vs. output relationship, as indicated by the broken line in Fig. 1B. In practice, however, the energy loss in the diode D₁₀ or D₁₁ distorts the ideal linear relationship, resulting in the curve shown by the dashed line in Fig. 1B. As shown Fig. 1C, this causes the small input gain (i.e. the gain when a drive pulse of small width is applied) to be lower than the theoretical value (indicated by the dashed line). The approximate energy loss which occurs in the two diodes due to absorption of reverse electromotive force will now be calculated. Fig. 2 is a diagram of an equivalent circuit of the switching drive circuit, and Fig. 3 shows waveforms of signals produced in response to a drive pulse (a). If it is assumed that no energy loss occurs in either of the diodes for absorption of reverse electromotive force, the discharge curve will follow the solid line (1) Fig. 3. In fact, however, the diodes cause energy losses as indicated by the solid line (2), and the discharge curve follows the dashed line (3). The energy that is actually consumed in a resistor R is represented by the hatched area W A , and the energy that is lost in the diodes due to absorption of reverse electromotive force is represented by the hatched area W B . The sum of the two areas W A and W B represents the theoretical energy consumption in resistor R (W O = W A + W B ) , and the efficiency η is expressed by W A /W B . Assuming a discharge current of i(t), a peak current of I L (constant) and the absence of energy loss in either of the diodes due to absorption of reverse electromotive force, the theoretical energy consumption in resistor R (W O = W A ) may be calculated as follows in consideration of the relation i(t) = I L ·exp(-R/L)t : Assuming that current the I is zero at time T z , that the peak current on the dashed line (3) in Fig. 5 is i p , and that the peak current caused by reverse electromotive force in the diodes is I D (a constant), W A , or the actual energy consumption in resistor R, may be calculated as follows: T Z may be calculated as follows by using the relation i p = (I L +I D )(exp(-R/L)T Z )-I D . Since i p = 0 at T Z , Substituting Eq. (3) into Eq. (2), Further, W B , or the energy loss in the diodes due to absorption of reverse electromotive force, may be determined as follows: The efficiency, η, can then be determined as follows: If the number of diodes used for absorbing reverse electromotive force is n and if each diode has a forward voltage drop of V F , the following relation may be written: ID = nV F /R. Subtituting this equation and I L = (E/R)(l-exp(-R L /L)T₀) into equation (6), the efficiency of a drive circuit using n diodes will be calculated as: = l-(2nV F /RI L )(l+nV F /RI L ·log e nV F /(RI L +nV F ) (7) If values of R = 10 ohms, E = 14.4 volts, L = 70 H, T₀ = 2 sec (20 sec x 10%), n = 2, and V F = 0.7 volts are substituted into equation (7), η is calculated as 0.606, which means that the efficiency of the drive circuit having these characteristics is 60.6%. If n = 1 and the other conditions are the same, the efficiency is increased to 74.7%. As will be understood from the foregoing explanation, diodes, which are essential in the switching drive circuit for absorbing reverse electromotive force, cause inevitable energy losses due to the reverse electromotive force they absorb. This energy loss is substantially constant and is small enough to be neglected if the drive pulse has a relatively great pulse width. On the other hand, if the pulse width is small, the relative proportion of the loss is increased and the efficiency of driving the load is lowered to such an extent that the desired drive energy cannot be attained. The configuration of another prior art PWM drive circuit is shown in Fig. 4B. Two triangular wave signals, a and b , which are in phase, are produced and fed to a comparator circuit 100 as upper and lower reference inputs, with one triangular signal a being biased to a higher d.c. level than the other triangle signal b . A drive signal c is fed to the circuit 100 as the comparison input. As a result, the circuit 100 produces a pair of pulse signals d and e which have pulse widths dependent on the signal level of the drive signal and which correspond to the two directions in which the load is to be driven. The load is driven by being switched on and off in response to the pair of pulse signals d and e . The waveforms of the signals a to e are shown in Fig. 4A. The PWM drive circuit employs a triangular generator circuit which conventionally has the configuration shown in Fig. 4C, wherein a rectangular wave signal having a predetermined period is converted to a triangular signal by an integrator 101 composed of an operational amplifier OP₁₀, resistors R₄₀ and R₄₁, and a capacitor C₁₀. It may be considered to fabricate this integrator-based triangular wave generator circuit in the form of an IC device, in which case two terminal pins P₁ and P₂ are necessary for making external connection to the capacitor C₁₀. However, one of the primary requirements for a circuit configuration that is adaptive for IC device fabrication is that it have a minimum number of terminal pins. Another difficulty with the conventional motor relates to the fact that it has a "dead zone" where it remains inactive until after the applied drive current exceeds a certain level. Due to the presence of this "dead zone", noise on the drive signal line will not cause the motor to start if the noise level is small. However, the power consumed by that current is still wasted. When the drive circuit is operated by the power supply connected to one end thereof, the reference level of the circuit, which is set to a value determined by division of the reference supply voltage by, for example, resistors, can vary from the desired level due to factors such as variations in the circuit elements and an offset in the signal level of the drive signal. This offset voltage causes a corresponding current to flow in the motor, even if the signal level is zero, and an extra power loss occurs as in the case of noise production. Such power loss should be eliminated since it inevitably leads to the use of a larger power unit. In addition, if the PWM drive circuit is used to drive motors in portable devices or those intended to be installed on vehicles, power consumption should be minimized since these devices use batteries as power sources. Smaller power consumption is also desirable in order to reduce the size and weight of these devices. A circuit configuration of the power drive stage wherein the two transistors on the power supply side are configured an npn transistors is shown in Fig. 5. In this circuit, a first pair of npn transistors Q₂₀ and Q₂₁ are connected in series with a load, for example, motor M, between a power supply V cc and ground, and a second pair of npn transistors Q₂₂ and Q₂₃ are also connected in series with the motor M. The circuit further includes pnp transistors Q₂₄ and Q₂₅ for driving the transistors Q₂₀ and Q₂₂ on the power supply side. When a forward driving pulse is supplied to transistor Q₂₄ through an inverter 20 and to transistor Q₂₁, both transistors Q₂₀ and Q₂₁ are turned on to provide the motor M with a drive current flowing in the direction indicated by a solid-line arrow in Fig. 5, thereby allowing the motor M to be driven in the forward direction. On the other hand, when a reverse driving pulse is supplied to transistor Q₂₅ through an inverter 21 and also to transistor Q₂₃, both transistors Q₂₂ and Q₂₃ are turned on to provide the motor M with a drive current flowing in the direction indicated by a dashed-line arrow, thereby driving the motor M in the reverse direction. One terminal of the motor M is connected to the power supply V cc via a diode D₁₀, while the other terminal of the motor is connected to V cc via another diode D₁₁. Both diodes serve to absorb reverse electromotive force. In this circuit, a higher efficiency is attained with a smaller number of diodes. Referring again to Fig. 5, transistors Q₂₀/Q₂₂ on the power supply side and transistors Q₂₁/Q₂₃ on the ground side are both subjected to duty-cycle control by a pulse signal. In addition, drive transistors Q₂₄ and Q₂₅ are provided in a stage preceding transistors Q₂₀ and Q₂₂ on the power supply side so to cause the transistors Q₂₀ and Q₂₂ to undergo a slower transition to the off state than transistors Q₂₁ and Q₂₃ on the ground side. If, on the other hand, in the forward driving mode, transistor Q₂₁ were to undergo a faster transistion to the off state than transistor Q₂₀, the energy stored in the coil of motor M would produce a reverse electromotive force which could cause current to flow through a closed loop including diode D₁₀, thereby resulting in a consumption of driving energy in the resistance of motor M. However, in this closed loop, the sum of the voltage drop of diode D₁₀ and the base-emitter voltage of transistor Q₂₀ can be regarded as V F in equation (7) above. Since this is equivalent to the presence of two diodes, a great energy loss occurs due to the reverse electromotive force, and the efficiency η is only about 60.6%. From US-A-4514666 a pulse width modulating circuit for a DC servo motor is known in which the fly-wheel current is used effectively to rapidly reduce the DC current when the circuit becomes non-driving or the driving polarity changes. A specific object of the invention is to provide a switching drive circuit which attains a higher efficiency in driving a load by compensating for the energy losses occurring in the diodes serving to absorb reverse electromotive force. Another object of the present invention is to provide a triangular wave generator circuit which can be fabricated in the form of an IC device with a reduced number of terminal pins. The PWM drive circuit of the present invention has a dead zone arranged so that no current will flow in the motor while it is in its dead zone. According to the present invention there is provided a switching drive circuit which has a unidirectional element for absorbing reverse electromotive force across a load and which drives the load by switching current flow through the load on and off in response to a predetermined pulse signal, a first transistor which supplies the load with a drive current for a forward direction in response to a first pulse signal, and a second transistor which supplies the load with a drive current for a reverse direction in response to a second pulse signal, said drive circuit driving said load by switching a current through said load on and off in response to said first and second pulse signals, characterized by: compensating circuit means receiving the predetermined pulse signal as an input, the compensating circuit means comprising means for generating a linearly varying signal on disappearance of the pulse signal, and means for comparing the varying signal with a reference signal to produce a drive pulse signal for the load, the drive pulse signal thus having a pulse width equivalent to a time interval from generation of the predetermined pulse signal and a point at a predetermined time after disappearance of the predetermined pulse signal so as to compensate for a predetermined pulse signal of small width and absorption of energy in the unidirectional element when the unidirectional element is absorbing the reverse electromotive force; and by the drive circuit further comprising: first switching means for shorting said first transistor between its base and emitter at a time of disappearance of said first pulse signal, and second switching means for shorting said second transistor between its base and emitter at a time of disappearance of said second pulse signal. An example of a circuit according to the present invention will now be described with reference to the accompanying drawings, in which:- Fig. 1A is circuit diagram showing the basic configuration of a conventional drive circuit; Fig. 1B is diagram showing the input/output characteristicS of the switching drive circuit of Fig. 1A; Fig. 1C is a diagram showing the gain characteristics of the switching drive circuit of Fig. 1A; Fig. 2 is a diagram showing an equivalent circuit of the switching drive circuit of Fig. 1A; Fig. 3 is a diagram showing a waveform which is generated in response to a drive pulse fed to the switching drive circuit of Fig. 1A; Fig. 4A is a waveform diagram showing signals in the switching drive circuit of Fig. 4B; Fig. 4B shows another example of a conventional switching drive circuit; Fig. 4C is a circuit diagram of a conventional triangular wave generating circuit; Fig. 5 is a schematic diagram showing still another example of a conventional switching drive circuit; Fig. 6 is a circuit diagram showing a first embodiment of the present invention; Fig. 7 is a waveform diagram which illustrates the operation of a triangular wave generator included in the circuit shown in Fig. 6; Fig. 8 is a waveform diagram which illustrates the generation of two pulse signals corresponding to the two directions in which a load is driven in a PWM operation; Figs. 9A and 9B are waveform diagrams which illustrate the operation for changing the ramp rate and peak value of a triangular wave in accordance with variations in the supply voltage; Fig. 10 is a waveform diagram which illustrates the operation of a circuit which prevents power transistors in a drive stage from being turned on simultaneously due to a delay time of each transistor; Fig. 11 is a diagram which illustrates the delay time of a transistor; Fig. 12 is a diagram which illustrates a gain deterioration caused by the energy loss occurring in reverse EMF absorbing diodes; Fig. 13 is a waveform diagram which illustrates the operation of a compensation circuit for compensating for the energy loss which occurs in the reverse EMF absorbing diodes; Fig. 14 is a diagram showing the input/output characteristics of the compensation circuit; Fig. 15 is a circuit diagram, similar to Fig. 6, showing a pulse-width modulation drive circuit of a second embodiment of the invention; Figs. 16 to 18 are waveform diagrams illustrating various operations of the circuit of Fig. 18; and Several preferred embodiments of the present invention will hereinafter be described in detail with reference to the drawings. Fig. 6 shows a first constant-current source 1 which is formed of a current-type Miller circuit composed of transistors Q₁ and Q₂ and resistors R₁ and R₂. This first constant-current source 1 is series-connected with a second constant-current source 2, which is formed of a current-type Miller circuit composed of parallel-connected transistors Q₃ and Q₄, transistor Q₅ connected to Q₃ and Q₄ at its base with a resistor R₃ inserted therebetween, and emitter resistances R₄ and R₅ for Q₃/Q₄ and Q₅, respectively. The second constant-current source 2 is designed to draw a curent 2I₀ which is twice the constant current I₀ supplied by the first current source 1. The common junction of the first and second constant-current sources 1 and 2, or the common-collector junction between transistors Q₂ and Q₃/Q₄, is connected to ground, serving as a reference potential point, and a charge-storing capacitor C₁ is connected between the common-collector junction and ground. The voltage across the capacitor C₁ is fed to a comparator circuit 3, which is composed of comparators, COMP₁ and COMP₂ and which serves as a monitor for the level of the terminal voltage, and is applied to the two comparison inputs of the circuit 3, namely, the inverting input of COMP₁ and the noninverting input of COMP₂. Upper and lower reference levels V U and V L for comparison by the circuit 3 are set to values determined by division of a reference source voltage V ref by four series-connected resistors R₅ to R₈. Another function of resistors R₅ to R₈ is to divide V ref approximately in half. The divided voltage is fed to an operational amplifier OP₁ which is configured as a voltage-follower circuit and which produces (1/2)V ref as its output. The two comparison outputs of the circut 3, namely, the outputs from COMP₁ and COMP₂, are fed to an R-S flip-flop 4 on its set (S) and reset (R) inputs, respectively. The Q output of the flip-flop (hereinafter abbreviated as FF) 4 is fed to a control circuit 5 which is composed of a transistor Q₆ and resistors R₉ and R₁₀ and which selectively activates and deactivates the second constant-current source 2. In this control circuit 5, transistor Q₆ turns on in response to the Q output of FF 4, and, as a result, transistors Q₃ and Q₄ are turned off to thereby deactivate the second constant-current source 2. The voltage across the emitter resistance R₅ in the second constant-current source 2 is fed to the inverting input of an operational amplifier OP₂ configured as a voltage-follower circuit. The reference level for comparison by OP₂ is set to a value determined by the division of V ref by resistors R₁₁ and R₁₂. The operational amplifier OP₂ provides a current value setting circuit 6 which uses the comparison output of the amplifier in setting the values of constant currents supplied from the first and second current sources 1 and 2. The voltage across capacitor C₁ is also fed to an operational amplifier OP₃ which is configured as a voltage-follower circuit and which produces a triangular-wave signal  a having a first phase. At the same time, the terminal voltage of C₁ is inverted in phase by an inverter 7 which is composed of an operational amplifier OP₄ and resistors R₁₃ and R₁₄ and which produces a triangular wave signal  b having a second phase opposite from the phase of the triangular-wave signal  a . Each of the two triangular-wave signals  a and  b is supplied with a d.c. bias of (1/2)V ref . The aforementioned components 1 to 7 form a triangular wave generator circuit 8 which produces two triangular-wave signals  a and  b having substantially equal peak values and which are opposite to each other in phase. The triangular wave generator circuit 8 includes the first constant-current source 1 providing a constant current output of I₀ and the second constant-current source 2 providing a constant current output of 2I₀, and is designed so as to produce a triangular wave by charging and discharging the capacitor C₁ with a constant current as a result of on/off control of the second constant-current source 2. This circuit 8 has the advantage that it can be formed in an IC device with only one additional input/output pin (terminal 8a in Fig. 6), namely, that for the capacitor C₁. The triangular wave signals  a and  b of opposite phases are supplied as the upper and lower reference inputs for comparison by a window comparison circuit 9 composed of comparators COMP₃ and COMP₄. More specifically,  a is fed to the inverting input of COMP₃ and  b to the inverting input of COMP₄. The comparison inputs of circuit 9, namely, the noninverting inputs of COMP₃ and COMP₄, are supplied with a drive signal for a load, specifically, motor M, through a resistor R₁₅. The noninverting input terminals of COMP₃ and COMP₄ are also fed with a reference source voltage V ref through a resistor R₁₆ (R₁₅ = R₁₆) . Since R₁₅ and R₁₆ have equal values of resistance, the drive signal will be biased by (1/2)V ref when it is fed to the comparison inputs of the window comparator 9. In other words, the signal reference level of the drive signal is set to 1/2(V ref ). As a result, both the circuit reference level of the triangular wave generator circuit 8, namely, the comparison reference level of the comparator circuit 3, and the d.c. bias level (signal reference level) of the drive signal are determined by resistance-based division of the same reference source voltage V ref . Therefore, even if variations occur in the source voltage, the relative signal levels of the triangular wave signals  a and  b and the drive signal are maintained constant, thereby ensuring consistent operation of the circuit 8. The comparison output of COMP₃ is fed to one input of each of an AND gate 10 and a NOR gate 11, whereas the comparison output of COMP₄ is fed to the other input of each gate. As a result, first and second pulse signals associated with the two directions of rotation of motor M are produced at the output terminals of AND gate 10 and NOR gate 11. The aforementioned drive signal is also fed to the noninverting input of a comparator COMP₅ through resistor R₁₅. The comparator COMP₅, which receives (1/2)V ref on its inverting input, forms a polarity determining circuit which determines the polarity of the drive signal with respect to the signal reference level. The decision output from COMP₅ is fed to the data (D) input of a D-FF 12, which receives at its trigger (T) input the Q output of R-S FF 4 in the triangular wave generator circuit 8. The Q output of D-FF 12 is fed to one input of an AND gate 13, and the Q output is fed to one input of an AND gate 14. The other input of AND gate 3 receives the output of AND gate 10 (i.e., the first pulse signal), and the other input of AND gate 14 receives the output of NOR gate 11 (i.e., the second pulse signal). The two AND gates 13 and 14 form a gate circuit which delivers only one of the first and second pulse signals as its output as determined by the Q and Q outputs of D-FF 12. The output pulses from AND gates 13 and 14 are supplied to compensation circuits 15 and 16 which serve to compensate for the energy loss caused by the reverse electromotive force applied to the diodes D₁ and D₂ in a motor drive circuit 18 (to be described in detail below). In the compensation circuit 15, the output pulse from AND gate 13 is fed to the base of a transistor Q₇ through a resistor R₁₇, the transistor Q₇ being connected in parallel with a capacitor C₂. When transistor Q₇ is turned on, the terminals of capacitor C₂ are shorted, causing the stored charge to be quickly discharged. A constant-current source I a in the compensation circuit 15 starts to charge the capacitor C₂ at the point of time when transistor Q₇ is turned off, or when the output pulse from the AND gate 13 ceases. The voltage across the capacitor C₂ is fed to the inverting input of a comparator COMP₆, which receives a reference voltage E₀ on its noninverting input and produces a high-level pulse signal when the voltage acrosS the capacitor C₂ is higher than E₀. As a consequence, the compensation circuit 15 produces a pulse signal which has a pulse of a substantially constant pulse width added to the output pulse from the AND gate 13. The compensation circuit 16 is configured in a manner similar to the compensation circuit 15; that is, it is composed of a resistor R₁₃, a transistor Q₈, a capacitor C₃, a constant-current source I b , and a comparator COMP₇, and it is operated in entirely the same manner as the compensation circuit 15. The output pulses from the compensation circuits 15 and 16 are supplied to a motor drive circuit 18 through a predrive circuit 17. In the motor drive circuit 18, motor M is connected between the common-collector junction of a pnp transistor Q₉ and an npn transistor Q₁₀ and the common-collector junction of a pnp transistor Q₁₁ and an npn transistor Q₁₂. The transistors Q₉, Q₁₀, Q₁₁ and Q₁₂ are power transistors. The emitters of transistors Q₉ and Q₁₁ are directly connected to a power source V cc , while their bases are connected to V cc via resistors R₁₉ and R₂₀. The emitters of transistors Q₀ and Q₁₂ are grounded, and their bases are grounded via resistors R₂₁ and R₂₂ and connected to their collectors via zener diodes ZD₁ and ZD₂, respectively. The two terminals of motor M are connected to V cc via diodes D₁ an D₂, respectively, which serve to absorb reverse electromotive force. In the predrive circuit 17, the pulse signal supplied from the compensation circuit 15 passes through a predrive stage, composed of resistors R₂₃ and R₂₄ and a transistor Q₁₃, and drives the power transistor Q₉. Further the pulse signal is inverted by an inverter 19, passes through a predrive stage, composed of resistors R₂₅, R₂₆ and R₂₇ and a transistor Q₁₄, and drives the power transistor Q₁₂. As a result, a current flows in the motor M in the direction indicated by a solid-line arrow in Fig. 6, causing the motor M to rotate in the forward direction. The pulse signal from the compensation circuit 15 is also supplied to a transistor Q₁₅ through an inverter 20, turning Q₁₅ on when the forward driving of the motor M ends. When Q₁₅ is turnd on, the power transistor Q₁₂ is shorted between its base and the emitter and is turned off substantially instantaneously. (The reason for the inclusion of this transistor Q₁₅ will be described in detail below.) The base of Q₁₅ is connected to the power source V cc via a resistor R₂₈. The pulse signal supplied from the compensation circuit 16 passes through a predrive stage composed of resistors R₂₉ and R₃₀ and a transistor Q₁₆, and drives the power transistor Q₁₁. Also, the pulse signal is inverted by an inverter 21, passes through a predrive stage composed of resistors R₃₁, R₃₂ and R₃₃ and a transistor Q₁₇, and drives the power transistor Q₁₀. As a result, a current flows in the motor M in the direction indicated by a dashed-line arrow in Fig. 6, driving the motor M in the reverse direction. The constant current from the compensation circuit 16 is also supplied to a transistor Q₁₈ through an inverter 22, turning Q₁₈ on when the reverse driving of the motor M ends. When Q₁₈ is turned on, the power transistor Q₁₀ is shorted between its base and emitter and is turned off substantially instantaneously. The base of Q₁₈ is connected to V cc via a resistor R₃₄. The operation of the PWM drive circuit in accordance with the present invention will now be described. First, the operation of the triangular wave generator circuit 8 will be described with reference to the waveform diagram of Fig. 7. When the second constant-current source 2 in the circuit 8 is in the deactivated state, in other words, when transistor Q₆ is on and transistors Q₃ and Q₄ are off, the constant current supplied from the first constant-current source 1 causes the capacitor C₁ to be charged at a constant rate as shown in Fig. 7, waveform (a). When the voltage across capacitor C₁ reaches the upper reference level V U for the comparator circuit 3, the comparator COMP₁ generates a low-level pulse (b), in response to which the Q output (d) of R-S FF 4 makes a transition to the low level. As a result, transistor Q₆ is turned off and the second constant-current source 2 is activated (i.e., transistors Q₃ and Q₄ are turned on) to draw a current twice the magnitude of the constant current supplied from the first constant-current source 1. This causes the charge stored on capacitor C₁ to be discharged at the same rate as the charging rate, as shown in waveform (a) of Fig. 7. When the voltage across capacitor C₁ reaches the lower reference level V L for the comparator circuit 3, the comparator COMP₂ generates a low-level pulse (c), in response to which the Q output (d) of R-S FF 4 makes a transition to the high level. As a result, transistor Q₆ is turned on and the second constant-current source 2 is deactivated, which allows capacitor C₁ to be recharged at a constant rate by the constant current supplied from the first constant-current source 1. As capacitor C₁ is repeatedly charged and discharged by the constant currents supplied from the first and second constant-current sources 1 and 2, the voltage across capacitor C₁ changes cyclically to form a triangular wave, as shown by the solid line of waveform (a) in Fig. 7. This triangular wave signal is fed to the operational amplifier OP₃, from which it is delivered as the triangular wave signal  a of the first phase. The output from OP₃ is also phase-inverted by the inverter 7 to produce the triangular wave signal  b of the second phase which, as shown by the dashed line in waveform (a) of Fig. 7, has the same peak value as  a and is opposite in phase to  a . The two triangular wave signals  a and  b are fed to the comparator circuit 9 as reference inputs. The comparison inputs of the circuit 9 are fed with the motor drive signal having a signal reference level of (1/2)V ref . If motor M is a spindle motor, such as may be used for rotating a compact disk, the drive signal is an error signal that is produced by comparing a reproduced sync signal from the disk with a reference sync signal. This error signal provides a basis for controlling the drive of the spindle motor. The drive circuit is then generally referred to as a spindle servo. The PWM operation of the drive circuit will hereunder be described with reference to Fig. 8 with respect to two cases where the signal level of the drive signal is higher and lower than the level of (1/2)V ref at which the two triangular signals  a and  b of opposite phases cross each other. First, if the signal level of the drive signal which is fed to the comparator circuit 9 is higher than the level of (1/2)V ref , as indicated by the one-short-and-one-long dashed line of waveform (a) in Fig. 8, the output (b) of the comparator COMP₃ makes a transition from the low to the high level at time t₁ when the signal level of the triangular wave signal  a of the first phase becomes lower than the signal level of the drive signal, and the resulting high level is maintained up to time t₄ when the signal level of  a exceeds that of the drive signal. On the other hand, the output (c) of comparator COMP₄ makes a transition from the high to the low level at time t₂ when the signal level of the triangular wave signal  b of the second phase exceeds the signal level of the drive signal, and makes a transition to the initial high level at time t₃ when the signal level of  b becomes lower than that of the drive signal. If, on the other hand, the signal level of the drive signal is lower than the level of (1/2)V ref , as indicated by the two-short-and-one-long dashed line of waveform (a) in Fig. 8, and if the drive signal level has the same absolute value as in the case described above, the output (d) of the comparator COMP₃ makes a transition from the low to the high level at the time t₂ when the signal level of the triangular wave signal  a of the first phase exceeds the signal level of the drive signal. The resulting high level is maintained to time t₃ when the signal level of  a exceeds that of the drive signal. The output (e) of the comparator COMP₄ makes a transition from the high to the low level at time t₁ when the signal level of the triangular wave signal  b of the second phase exceeds the signal level of the drive signal, and makes a transition to an initial high level at time t₄ when the signal level of  b becomes lower than that of the drive signal. The outputs from comparators COMP₃ and COMP₄ provide the two inputs for both AND gate 10 and NOR gate 11. When the two inputs to AND gate 10 are at a high level, in other words, when the signal level of the drive signal is higher than (1/2)V ref , the AND gate 10 produces a high-level Pulse (f). When the two inputs to NOR gate 11 are at the low level, in other words, when the signal level of the drive signal is lower than (1/2)V ref , the NOR gate 11 produces a high-level pulse (g). That is, the AND gate 10 and NOR gate 11 produce pulse signals as shown in waveforms (f) and (g), respectively, as outputs associated with the two rotational directions of the motor M. The foregoing description assumes that the drive signal has a constant signal level and that the pulse signals (f) and (g) have constant pulse widths. It may, however, be readily understood that the pulse width of each of the pulse signals (f) and (g) is variable depending upon the signal level of the drive signal. As described above, in accordance with the present invention, two triangular wave signals  a and  b which have equal peak values and are opposite in phase are produced, and the linear portion of each of these triangular wave signals is used to perform the PWM operation. This eliminates any degradation of the linearity of the input vs. output characteristics at small drive signal levels, even if ringing occurs at the apex of a cycle of the triangular wave or if it has a flattened end. If variations occur in the reference source voltage V ref , the pulse width of the pulse signal generated by PWM will change, and the drive power generated in response to this pulse signal will also change in accordance with the variations in the source voltage. If the pulse signal has a pulse width of T₀ at a certain drive signal level as shown in Fig. 9A, the drive power generated in response to this pulse signal is defined by the product of T₀ and the drive voltage V D (with respect to the reference source voltage V ref ). Therefore, if the drive voltage V D is reduced by half as a result of variation of the source voltage, the drive power is also reduced to half of its initial level, as indicated by the hatched area in Fig. 9A. Further, in the triangular wave generator circuit 8, the comparison reference levels for the current value setting circuit 6, which serves to set the values of the constant currents supplied from the first and second constant-current sources 1 and 2, are determined by division of the reference source voltage V ref by resistors R₁₁ and R₁₂, and hence will follow the variations in the source voltage. Because of this feature, the current value setting circuit 6 is capable of controlling the values of the constant currents from the first and second constant-current sources 1 and 2 in accordance with the variation in the source voltage, thereby making the ramp of the triangular wave change as shown in Fig. 9B. In addition, the upper and lower reference levels V U and V L used for comparison by the circut 3 are set to values which are determined by division of V ref by resistors R₅ to R₈. Therefore, if V ref is reduced by half, the upper and lower reference levels V U and V L are also reduced by half, with the result that the peak value V P of the triangular wave is reduced to half the initial value (i.e., the value attained before the variations in the source voltage), as shown in Fig. 9B. In other words, if the ramp rate of the triangular wave is set such that the repetition rate of the signal is the same before and after any variation in the source voltage, a pulse signal will be formed whose pulse width is 2T₀, twice the width attained before the variation in the source voltage. Therefore, even if a variation occurs in the source voltage and the drive voltage V D is reduced by half, the drive power generated in response to the pulse signal will remain unchanged. In other words, the triangular wave generator circuit 8 controls the peak value and ramp rate of the triangular wave in accordance with any variation that is introduced into the source voltage, and, as a result of this control action, the drive power generated in response to the pulse signal is maintained constant, irrespective of the variations in the reference source voltage V ref . In this connection, it should be mentioned that the ramp rate of the triangular wave is determined by the values of the constant currents supplied from the first and second constant-current sources 1 and 2 and by the capacitance of the capacitor C₁. Returning now to Fig. 6, if the signal level of the drive signal changes as shown by the one-short-and-one-long dashed line in Fig. 10, waveform (a), two pulse signals (b) and (c) having pulse widths corresponding to the polarity and signal level of the drive signal are produced from the outputs of AND gate 10 and NOR gate 11. Signal (b) is fed to one input of the AND gate 13 while signal (c) is fed to one input of the AND gate 14. The drive signal is also fed to the comparison input of comparator COMP₅ for the purpose of determining the polarity of the drive signal with respect to the signal reference level (1/2)V ref . The comparison output (d) of COMP₅ is fed to the data (D) input of D-FF 12, which is triggered by the Q output (e) of R-S FF 4 in the triangular wave generator circuit 8, and produces Q and Q outputs (f) and (g), respectively, at the timing determined by the fall of the Q output (e) of R-S FF 4. The Q and Q outputs (f) and (g), respectively, produce gate control signals which are supplied to the AND gates 13 and 14, respectively. In the embodiment described above, the Q output (e) of R-S FF 4 is directly fed to the trigger input of D-FF 12. Alternatively, the Q output (e) of R-S FF 4 may be fed to the trigger input T of D-FF 12 through a pulse generator that generates pulses at the rise and fall of the Q output (e). This modification reduces the cycle period of polarity decision by half and doubles the resolution. The Q and Q outputs (f) and (g), respectively, of D-FF 12 serve as control signals for determining the direction in which the motor M is driven. If the drive signal is at the low signal level and a pulse signal for effecting motor driving in the reverse direction is generated from the NOR gate 11 momentarily when the polarity of the drive signal changes from positive to negative (as indicated by the first pulse (c) in Fig. 10), at the point of time when such reverse driving pulse signal is generated, the Q output (g) is at low level and the AND gate 14 is operated so as to inhibit the outputting of the Q signal. The reason for performing this inhibitory operation will be described below. It is assumed that the drive signal is at the low signal level and also that a pulse of the signal for effecting motor driving in the reverse direction is generated from the NOR gate 11 momentarily when the drive signal (c) changes its polarity from positive to negative as shown in Fig. 10. In the motor drive circuit 18, transistors Q₉ and Q₁₂ are thus turned on in response to the pulse signal (b) shown in Fig. 10 to drive the motor M to rotate in the forward direction. However, when a pulse of the signal (c) for driving in the reverse direction is generated, transistors Q₉ and Q₁₂ are turned off whereas transistors Q₁₁ and Q₁₀ are turned on to drive the motor M to rotate in the reverse direction. As shown in Fig. 11, a transistor has a capacitance C₀ between its base and emitter. When a drive pulse (a) is applied to a transistor in the conducting (on) state, the transistor will be turned off only when a certain delay time (t off ) lapses after the pulse (a) ceases to exist. If a pulse signal (c) for driving in the reverse direction is generated as shown in Fig. 10, transistors Q₉ and Q₁₂ will theoretically be turned off while transistors Q₁₁ and Q₁₀ are turned on. In fact, however, the delay time t off prevents transistor Q₁₂ from being turned off instantly, and hence both transistors Q₁₂ and Q₁₁ will remain conducting temporarily. As a result, a large current will flow in transistors Q₁₁ and Q₁₂ which could cause their failure. The PWM drive circuit of the present invention includes AND gates 13 and 14 which are controlled in response to the polarity of the drive signal with respect to the signal reference level. If a pulse signal (c) for driving in the reverse direction is generated as shown in Fig. 10, AND gate 14 will inhibit the outputting of the pulse signal in response to the Q output (g) of D-FF 12, which prevents transistor Q₁₂ from being turned on simultaneously with transistor Q₁₁. An additional feature of the PWM drive circuit of the present invention is that transistors Q₁₅ and Q₁₈ are provided for the purpose of reducing the delay time t off of each of the power transistors Q₁₂ and Q₁₀. The transistors Q₁₅ and Q₁₈ are turned on momentarily in response to the disappearance of the drive pulse that was applied to power transistors Q₁₂ and Q₁₀, and the transistors Q₁₂ and Q₁₀ are shorted between their base and emitter, thereby shortening the delay time t off of each of the power transistors Q₁₂ and Q₁₀. Transistors typically have a delay time (t off ) ranging from about 1 to 2 µsec, but this can be reduced by about one order of magnitude to approximately 100 nsec because of the inclusion of transistors Q₁₅ and Q₁₈. In the circuit shown in Fig. 6, the first and second pulse signals from the AND gates 13 and 14 associated with the two respective rotational directions in which the motor M is driven, are supplied to the compensation circuits 15 and 16, respectively. The compensation circuits 15 and 16 serve to compensate for the energy loss which occurs in reverse EMF absorbing diodes D₁ and D₂ in the motor drive circuit 18. The energy loss occurring in the diodes D₁ and D₂ is substantially constant and is negligible if the pulse signal has a large pulse width, but the proportion of this energy loss increases as the pulse width decreases. As a result, the gain will drop in the region where the pulse signal has a small pulse width, as indicated by the dashed line in Fig. 12. In other words, the energy loss occurring in the reverse EMF absorbing diodes D₁ and D₂ needs to be compensated when the pulse signal has a small pulse width. The operation of the compensation circuit 15 will hereunder be described with reference to the waveform diagram of Fig. 13. Capacitor C₂ has been charged with a constant current by the constant-current source I a . When transistor Q7 turns on in response to an input pulse (a), the charge stored in C₂ is discharged momentarily. When the input pulse (a) ceases the capacitor C₂ is recharged at a constant rate. Therefore, the voltage (b) across C₂ changes as shown in Fig. 13. The terminal voltage (b) is compared with a reference voltage E₀ by comparator COMP₇, and, as a result, the output terminal of COMP₇ is provided with a pulse signal (c) having a pulse width which is equivalent to the time interval from the generation of the input pulse (a) to the time when a given time T a lapses after the disappearance of the input pulse. In other words, a given pulse width T a is added to the input pulse (a), and the energy corresponding to the added pulse width T a serves to compensate for the energy loss occurring in the reverse EMF absorbing diodes D₁ and D₂. Fig. 14 shows the input/output characteristics of the compensation circuits 15 and 16, or the relationship between the pulse width of an input pulse and the pulse width added thereto. In region (1) of the pulse width of an input pulse where the voltage across capacitor C₂ is higher than the reference voltage E₀ for comparator COMP₇, there is no addition of pulse width; in region (2) where the voltage across C₂ drops below E₀ and approaches zero, the added pulse width increasing linearly; and in region (3) where the voltage across C₂ levels off at zero, a fixed pulse width is added to the input pulse. In regions (1) and (2) where the input pulse has a very small pulse width, no addition of pulse width occurs or the added pulse width increases linearly due to the fact that, in actual operation, the input pulse rises and falls with a gradual slope rather than sharply, with the result that the gain attained in region (2) is improved, as shown by the solid line in Fig. 12. The description of the foregoing embodiments assumes that the present invention is applied to a circuit for driving a spindle motor which drives a compact disk or the like. It should, however, be noted that the present invention is also applicable to various drive circuits such as a circuit for driving a pickup driving carriage motor, a circuit for driving a focus actuator which controls the focusing of light used in reading information with a pickup, and a circuit for driving a tracking actuator which controls the tracking of such light. The application of the present invention is by no means limited to compact disk players, and it may be applied to various other circuits used in driving loads in a variety of machines. Another preferred embodiment of the invention will now be explained with reference to Figs. 15 to 18. Fig. 15 is a schematic diagram of a PWM drive circuit of this embodiment of the invention. As the embodiment of Fig. 15 is similar in many respects to that of Fig. 6, the discussion below relates primarily only to the differences between the two. The triangular signals  a and  b of opposite phases are supplied to first and second comparator circuits 9A and 9B. The first comparator circuit 9A is composed of a pair of comparators COMP₃ and COMP₄ which receive the signals  a and  b on their noninverting inputs. The second comparator circuit 9B is composed of a pair of comparators COMP₅ and COMP₆ which receive the signals  a and  b on their inverting inputs. The inverting inputs of COMP₃ and COMP₄ and the noninverting inputs of COMP₅ and COMP₆ are fed with a drive signal for motor M which has its signal reference level shifted by resistors R₁₅ to R₁₈ connected in series between an input terminal P₁ and a reference source (V ref ) terminal P₂. The resistors R₁₅ and R₁₈ have the same value of resistance, as do the resistors R₁₆ and R₁₇. The values of resistors R₁₅ and R₁₈ are comparatively larger than those of R₁₆ and R₁₇. As a result of these resistance values, the drive signal fed to the inverting input of each of the comparators COMP₃ and COMP₄ is shifted from the center level of each triangle signal ((1/2)V ref ) by a predetermined positive level (+ΔV), while the drive signal fed to the inverting input of each of the comparators COMP₅ and COMP₆ is shifted from the center level ((1/2)V ref ) by a predetermined negative level (-ΔV). The outputs of comparators COMP₃ and COMP₄ are fed to the two inputs of an AND gate 10, while the outputs of comparators COMP₅ and COMP₆ are fed to the two inputs of an AND gate 11. As a result, first and second pulse signals corresponding to the two directions in which the motor M is to be driven are produced at the output terminals of AND gates 10 and 11. The aforementioned drive signal is also fed to the noninverting input of a comparator COMP₇ after it has been provided with the signal reference level of (1/2)V ref by means of resistors R₁₅ to R₁₈. The comparator COMP₇ receives (1/2)V ref at its noninverting input and determines the polarity of the drive signal with reference to the signal reference level. The decision output from COMP₇ is fed to the data (D) input of a D-FF 12, which receives at its trigger (T) input the Q output of R-S FF 4 in the triangular generator circuit 8. The Q output of D-FE 12 is fed to one input of an AND gate 13, and the Q output is fed to one input of an AND gate 14. The other input of AND gate 13 receives the output of AND gate 10 (i.e., first pulse signal), and the other input of AND gate 14 receives the output of AND gate 11 (i.e., second pulse signal). The two AND gates 13 and 14 form a gate circuit which delivers only one of the first and second pulse signals as its output as determined by the state of the Q and Q outputs of D-FF 12. The output pulses from the AND gates 13 and 14 are supplied to compensation circuits 15 and 16 which serve to compensate for the energy loss occurring in reverse EMF absorbing diodes D₁ and D₂ in a motor drive circuit 18. The compensation circuits 15 and 16 output pulse signals composed of pulses of a substantially constant pulse width added to the output pulses from AND gates 13 and 14. The output pulses from compensation circuits 15 and 16 are supplied to the motor drive circuit 18 through a predrive circuit 17. In the motor drive circuit 18, motor M is connected between the common-collector junction of a pnp transistor Q₉ and an npn transistor Q₁₀ and the common-collector junction of a pnp transistor Q₁₁ and an npn transistor Q₁₂. The transistors Q₉, Q₁₀, Q₁₁ and Q₁₂ are power transistors. The emitters of transistors Q₉ and Q₁₁ are directly connected to a power supply V cc , while their bases are connected to V cc via resistors R₁₉ and R₂₀. The emitters of transistors Q₁₀ and Q₁₂ are grounded; and their bases are grounded via resistors R₂₁ and R₂₂ and connected to their collectors via zener diodes ZD₁ and ZD₂, respectively. The terminals of motor M are connected to V cc via diodes D₁ and D₂ which serve to absorb reverse electromotive force. In the predrive circuit 17, the pulse signal supplied from the compensation circuit 16 passes through a predrive stage composed of resistors R₂₃ and R₂₄ and a transistor Q₁₃ and drives the power transistor Q₉. At the same time, the pulse signal is inverted by an inverter 19, passes through a predrive stage composed of resistors R₂₅ to R₂₇ and a transistor Q₁₄, and drives the power transistor Q₁₂. As a result, a current flows in the motor M in the direction indicated by a solid-line arrow (Fig. 18) and causes the motor M to rotate in the forward direction. The pulse signal from the compensation circuit 16 is also supplied to a transistor Q₁₅ through an inverter 20 and turns Q₁₅ on when the forward driving of the motor M ends. When Q₁₅ is turned on, the power transistor Q₁₂ is shorted between its base and emitter, turning it off substantially instantaneously. The base of Q₁₅ is connected to the power supply V cc via a resistor R₂₈. The pulse signal supplied from the compensation circuit 15 passes through a predrive stage composed of resistors R₂₉ and R₃₀ and a transistor Q₁₆, and drives the power transistor Q₁₁. At the same time, the pulse signal is inverted by an inverter 21, passes through a predrive stage composed of resistors R₃₁ to R₃₃ and a transistor Q₁₇, and drives the power transistor Q₁₀. As a result, a current flows in the motor M in the direction indicated by the dashed-line arrow (Fig. 15), and the motor M is driven to rotate in the reverse direction. The constant current from the compensation circuit 15 is also supplied to a transistor Q₁₈ through an inverter 22 and turns Q₁₈ on when the reverse driving of the motor M ends. When Q₁₈ is turned on, the power transistor Q₁₀ is shorted between its base and emitter, and is turned off substantially instantaneously. The base of Q₁₈ is connected to V cc via a resistor R₃₄. The comparison inputs of the first and second comparison circuits 9A and 9B are fed with motor drive signals having signal reference levels of (1/2)V ref + ΔV and (1/2)V ref - ΔV . If motor M is a spindle motor for rotating a spindle disk, each of the drive signals may be an error signal produced by comparing a reproduced sync signal from the disk with a reference sync signal. This error signal provides a basis for controlling the driving of the spindle motor. The PWM operation of the drive circuit will hereunder be described with reference to the waveform diagrams of Figs. 16 to 18 with respect to three cases: where the signal level of the drive signal before level shifting is zero, greater than a predetermined level by more than + ΔV, and greater than the predetermined level by less than + ΔV. Figs. 16 to 18 are waveform diagrams which illustrate the operation of the PWM circuit shown in Fig. 15; (a) to (f) show the waveforms of signals (a) to (f) noted in Fig. 15. Suppose first the case where the drive signal level is zero (see Fig. 16). The signal level of the drive signal fed to the first comparator circuit 9A is level-shifted to (1/2)V ref + ΔV as indicated by the one-short-and-one-long dashed line, and the output (a) of comparator COMP₃ makes a transition from the high to the low level at time t₁ when the signal level of the triangular signal  a of the first phase becomes lower than the signal level of the level-shifted drive signal. The resulting low level is maintained up to time t₄ when the signal level of  a exceeds that of the drive signal. On the other hand, the output (b) of comparator COMP₄ makes a transition from the low to the high level at time t₂ when the signal level of the triangular signal  b of the second phase exceeds the signal level of the drive signal, and makes a transition to the initial low level at time t₃ when the signal level of  b becomes lower than that of the drive signal. However, the output (c) of AND gate 10, to the inputs of which are supplied the outputs (a) and (b), is at the low level at this point of time, and no output pulse signal will be produced by the AND gate 10. With further reference to Fig. 16, the signal level of the drive signal fed to the second comparator circuit 9B is level-shifted to (1/2)V ref - ΔV as indicated by the two-short-and-one-long dashed line, and the output (d) of comparator COMP₅ makes a transition from the high to the low level at time t₁ when the signal level of the triangular signal  b of the second phase becomes higher than the signal level of the level-shifted drive signal. The resulting low level is maintained up to time t₄ when the signal level of  b becomes lower than the signal level of the drive signal. On the other hand, the output (e) of comparator COMP₆ makes a transition from the low to the high level at time t₂ when the signal level of the triangular signal  a of the first phase becomes lower than the signal level of the drive signal, and makes a transition to the initial low level at time t₃ when the signal level of  a exceeds that of the drive signal. However, the output (f) of AND gate 11, to the inputs of which are fed the outputs (d) and (e), is at the low level at this point of time, and no output pulse signal will be produced by the AND gate 11. The case will now be considered where the drive signal level is greater than zero (ground level) by more than + ΔV (ΔV + α ) , with reference being made to the waveform diagram of Fig. 17. In this case, the signal level of the drive signal fed to the first comparator circuit 9A is level-shifted to (1/2)V ref + 2 ΔV + α as indicated by the one-short-and-one-long dashed line, and the output (a) of comparator COMP₃ makes a transition from the high to the low level at time t₁ when the signal level of the triangular signal  a of the first phase becomes lower than the signal level of the level-shifted drive signal, and the resulting low level is maintained up to time t₃ when the signal level of  a exceeds that of the drive signal. On the other hand, the output (b) of comparator COMP₄ makes a transition from the low to the high level at time t₄ when the signal level of the triangular signal  b of the second phase exceeds the signal level of the drive signal, and makes a transition to the initial low level at time t₅ when the signal level of  b becomes lower than that of the drive signal. However, the output (c) of AND gate 10, to the inputs of which are fed the outputs (a) and (b), is at the low level at this point of time and no output pulse signal will be produced by the AND gate 10. With further reference to Fig. 17, the signal level of the drive signal fed to the second comparator circuit 9B is level-shifted to (1/2)V ref + α as indicated by the two-short-and-one-long dashed line, and the output (d) of comparator COMP₅ makes a transition from the high to the low level at time t₃ when the signal level of the triangular signal  b of the second phase becomes higher than the signal level of the level-shifted drive signal. The resulting low level is maintained up to time t₆ when the signal level of  b becomes lower than the signal level of the drive signal. On the other hand, the output (e) of comparator COMP₆ makes a transition from the low to the high level at time t₇ when the signal level of the triangular signal  a of the first phase becomes lower than the signal level of the drive signal, and makes a transition to the initial low level at time t₇ when the signal level of  a exceeds that of the drive signal. As a consequence, the output (f) of AND gate 11, to the inputs of which are fed the outputs (d) and (e), attains a high level between t₂ and t₃ and also between t₆ and t₇, during which intervals the AND gate 11 produces an output pulse signal for forward driving. Finally, the case will be considered where the drive signal level is greater than zero by less than + ΔV (ΔV - α ) , with reference being made to the waveform diagram of Fig. 18. In this case, the signal level of the drive signal fed to the first comparator circuit 9A is level-shifted to (1/2)V ref - α as indicated by the one-short-and-one-long dashed line, and the output (a) of comparator COMP₃ makes a transition from the high to the low level at time t₁ when the signal level of the triangular signal  a of the first phase becomes lower than the signal level of the level-shifted drive signal. The resulting low level is maintained up to time t₄ when the signal level of  a exceeds that of the drive signal. On the other hand, the output (b) of comparator COMP₄ makes a transition from the high to the low level at time t₅ when the signal level of the triangular signal  b of the second phase becomes lower than the signal level of the drive signal, and makes a transition to the initial high level at time t₈ when the signal level of  b exceeds that of the drive signal. As a result, the output (c) of AND gate 10, to the inputs of which are supplied the outputs (a) and (b), attains a high level between t₄ and t₅ and also between t₈ and t₁, during which intervals the AND gate 10 produces an output pulse signal for reverse driving. Further referring to Fig. 18, the signal level of the drive signal fed to the second comparator circuit 9B is level-shifted to (1/2)V ref - ΔV - α as indicated by the two-short-and-one-long dashed line, and the output (d) of comparator COMP₅ makes a transition from the low to the high level at t₆ when the signal level of the triangular singal  b of the second phase becomes lower than the signal level of the level-shifted drive signal. The resulting low level is maintained up to time t₇ when the signal level of  b becomes higher than the signal level of the drive signal. On the other hand, the output (e) of comparator COMP₆ makes a transition from the low to the high level at time t₂ when the signal level of the triangular signal  a of the first phase becomes lower than the signal level of the drive signal, and makes a transition to the initial low level at t₃ when the signal level of  a exceeds that of the drive signal. However, the output (f) of AND gate 11, to the inputs of which are fed the outputs (d) and (e), is at the low level at this point of time and no output pulse signal is produced by the AND gate 11. As is apparent from the foregoing description, if the signal level of the drive signal before level shifting is within the range of zero ±ΔV, neither forward nor reverse driving pulses will be generated by the drive circuit of this embodiment of the present invention. In other words, the drive circuit of this embodiment is configured so that it is totally insensitive to any drive signal whose level is within the range of zero ±ΔV. Therefore, if the range of ±ΔV is appropriately determined to correspond to the dead zone of motor M, no current will flow in the motor M when it is in its dead zone and any accompanying power loss is eliminated. Referring back to Fig. 18, two pulse signals which correspond to the two different directions of motor driving are produced from AND gates 10 and 11, and the output from the AND gate 10 is fed to one input of AND gate 13, while the output from the AND gate 11 is fed to one input of AND gate 14. The drive signal is also fed to the comparison input of comparator COMP₇ for the purpose of determining the polarity of the drive signal with respect to the signal reference level (1/2)V ref . The comparison output of COMP₇ is fed to the data input of D-FF 12, which is triggered by the Q output of R-S FF 4 and produces Q and Q outputs at a timing determined by the falling of the Q output of R-S FF 4. The Q and Q outputs thus produced form gate control signals which are supplied to the AND gates 14 and 13. The first or second pulse signal produced by the AND gates 13 and 14, corresponding to the respective two directions in which the motor M is to be driven, is supplied to motor drive circuit 18 through compensation circuits 15 and 16 and through predrive circuit 17, thus selectively causing the motor M to rotate in either the forward or reverse direction. As described above, the switching drive circuit of the invention is configured so that the pulse width of an input pulse signal is increased by a predetermined degree to produce a pulse signal by which a load is driven. By employing the circuit configuration of the invention, the energy loss occuring in reverse EMF absorbing diodes (unidirectional elements) is compensated, with the result that the linearity of the input/output characteristics of the drive circuit while fed with a drive signal of a small pulse width are sufficiently improved to attain a higher efficiency in driving the load. Moreover, as the switching drive circuit of the present invention incorporates switching means for shorting the transistors in the power drive stage between their base and emitter momentarily when the applied drive pulses cease, the transistors are allowed to become fully off momentarily after the disappearance of the drive pulses, and thus the possibility of simultaneous turning on the transistors in the power drive stage because of the delay time (t off ) of each transistor can be effectively eliminated. In addition, the polarity of the drive signal is determined with respect to the signal reference level, and only a drive pulse that corresponds to a certain direction of driving, determined as a result of the polarity determination, is supplied to the power drive stage. As a result of this circuit design, the possibility of simultaneous turning on of transistors in the power drive stage because of the delay time (t off ) of these transistors is effectively eliminated. Also, a drive pulse for one direction is supplied to the power drive stage after it is delayed by a predetermined time, while the supply of a drive pulse for the other direction to the power drive stage is inhibited during the time interval between the generation of the first pulse signal and a point a predetermined time after the disappearance of the first pulse signal. As a result, the possibility of simultaneous turning on of the transistors in the power drive stage because of the delay time (t off ) of these transistors in effectively eliminated. Moreover, because the drive circuit of the present invention is configured so that by controlling the ramp rate and peak value of the triangular signals in accordance with variations in the supply voltage, the pulse width of the drive pulse is properly adjusted for the same level of drive signal. With this arrangement, the drive power is made constant for the same drive signal level, even if variations occur in the supply voltage, and, as a result, consistent circuit operation is ensured regardless of any variations in the supply voltage. Still further, in the PMW drive circuit of the present invention configured so that only the linear portion of a triangular signal is used for producing a load-driving pulse signal, the circuit is totally insensitive to any ringing or flattening that may occur at the apex of individual peaks of the triangular wave, and hence provides an input vs. output characteristic of improved linearity, even when the drive signal has a small signal level. Further, as described in the foregoing, the triangular wave generator circuit of the present invention includes a first constant-current source which has a constant current rating twice the value for the first constant-current source, and is configured so that it produces a triangular signal by charging and discharging a capacitor (charge storing means) at a constant current rate with on-off control being performed on the second constant-current source. In this circuit, only one end of the capacitor need be connected to a reference potential point, and hence the overall circuit can be fabricated as an IC device with the capacitor being externally connected at only one terminal pin, rather than the two pins required in the prior art circuit. Finally, as also described above, the power drive stage of a PWM drive circuit of the embodiment of the present invention which includes npn transistors on the power supply side is configured so that a unidirectional element for absorbing reverse electromotive force is connected between each terminal of the load and ground, and the transistors on the power supply side are driven by a pulse signal while those on the ground side are driven by a signal indicating the direction in which the load is to be driven. Due to this configuration, the PWM drive circuit of the present invention attains a higher efficiency in driving the load by reducing the energy loss occurring in the diodes (unidirectional elements) serving to absorb reverse electromotive force.

Claims: A switching drive circuit which has a unidirectional element (D₁,D₂) for absorbing reverse electromotive force across a load (M) and which drives the load by switching current flow through the load on and off in response to a predetermined pulse signal, a first transistor (Q₉,Q₁₂) which supplies the load (M) with a drive current for a forward direction in response to a first pulse signal (b,Fig.10), and a second transistor (Q₁₁,Q₁₀) which supplies the load with a drive current for a reverse direction in response to a second pulse signal (c,Fig.10), said drive circuit driving said load by switching a current through said load on and off in response to said first and second pulse signals, characterized by: compensating circuit means (15,16) receiving the predetermined pulse signal (a,Fig.13) as an input, the compensating circuit means comprising means (C₂,C₃) for generating a linearly varying signal (b,Fig.13) on disappearance of the pulse signal, and means (COMP₆,COMP₇) for comparing the varying signal with a reference signal (E₀) to produce a drive pulse signal (c,Fig.13) for the load (M), the drive pulse signal thus having a pulse width equivalent to a time interval from generation of the predetermined pulse signal (a,Fig.13) and a point at a predetermined time (T a ) after disappearance of the predetermined pulse signal so as to compensate for a predetermined pulse signal of small width and absorption of energy in the unidirectional element (D₁,D₂) when the unidirectional element is absorbing the reverse electromotive force; and by the drive circuit further comprising: first switching means (Q₁₅) for shorting said first transistor (Q₁₂) between its base and emitter at a time of disappearance of said first pulse signal, and second switching means (Q₁₈) for shorting said second transistor (Q₁₀) between its base and emitter at a time of disappearance of said second pulse signal. A switching drive circuit according to claim 1, further including a triangular wave generating means (8) for generating two triangular signals ( a , b ) having substantially equal peak values (V U ,V L ) which are opposite to each other in phase, a comparator circuit (9) receiving said two triangular signals of opposite phases as upper and lower reference input signals, respectively, and receiving a drive signal as a comparison input, first gate means (10,11) for producing first and second pulse signals ((f),(g);Fig.8) in response to an output of said comparator circuit, said first and second pulse signals corresponding to two directions in which said load is to be driven, polarity determining means (COMP₅,12) for determining a polarity of said drive signal with respect to a signal reference level, and second gate means (13,14) for outputting only one of said first and second pulse signals in response to a result of decision by said polarity determining means, the drive circuit (18) driving the load in response to an output of said second gate means. A switching drive circuit according to claim 2, wherein the triangular wave generating means includes a first constant-current source (1), a second constant-current source (2) connected in series with said first constant-current source and producing twice the amount of current as that produced by said first constant-current source; and including charge storing means (C₁) connected between a common junction (8) of said first and second constant-current sources and a reference potential point; a second comparator circuit (3) for monitoring an output level of said charge-storing means; control means (5) for selectively activating and deactivating said second constant-current source in response to an output of said second comparator circuit; current value setting means (R₅-R₈) for setting amounts of constant current produced by said first and second constant-current sources, said current value setting means controlling said amounts of constant current of said first and second constant-current sources in accordance with a variation in a supply voltage (V ref ), and means (R₅-R₈) for providing as a comparison reference level for said second comparator circuit a voltage determined by division of said supply voltage, said two triangular signals of opposite phases being outputted in response to an output signal from said charge-storing means. A switching drive circuit according to claim 1, including a triangular wave generating means (8) for generating two triangular signals ( a , b ) having substantially equal peak values and which are opposite to each other in phase, and a comparator circuit (9) receiving said two triangular signals of opposite phases as upper and lower reference inputs, respectively and said drive signal as a comparison input, said drive circuit producing said pulse signal in response to an output of said comparator circuit. A switching drive circuit according to claim 1, including a triangular wave generator circuit comprising: a first constant-current source (1), a second constant-current source (2) connected in series with said first constant-current source and supplying twice the amount of current as said first constant-current source, charge-storing means (C₁) connected between a common junction (8a) of said first and second constant-current sources and a reference potential point, a comparator circuit (3) for monitoring an output level of said charge-storing means, and control means (5) for activating said second constant - current source in response to an output of said comparator circuit, an output signal of said charge-storing means being produced as a triangular signal. A switching drive circuit according to claim 4, wherein a circuit reference level of said triangular wave generator circuit and a d.c. bias level of said drive signal are set to values determined by division of said supply voltage. A switching drive circuit according to claim 1, including a triangular wave generating means (8) for generating two triangular signals ( a , b ) which have substantially equal peak values and which are opposite to each other in phase, a first comparator circuit comprising pair of comparators (COMP₃,COMP₄;Fig.18) receiving said triangular signals of opposite phases on their non-inverting inputs and said drive signal shifted from a centre level of each triangular signal by a predetermined positive level on their inverting inputs, and a second comparator circuit comprising a pair of comparators (COMP₅,COMP₆;Fig.18) receiving said triangular signals of opposite phases on their inverting inputs and said drive signal shifted from a centre level of each triangular signal by a predetermined negative level on their non-inverting inputs, the drive circuit producing the pulse signal in response to outputs of said first and second comparator circuits. A switching drive circuit according to any of claims 1 to 7, wherein said first and second unidirectional elements (D₁,D₂) are Schottky diodes.

Inventors: ABE HIROYUKI PIONEER WORKS COR, AOYAGI YOSHIO PIONEER WORKS CO, HAENO AKIRA PIONEER WORKS CORP, IIJIMA TAKAYUKI PIONEER WORKS, KIMURA TOSHIYUKI PIONEER WORKS, KOBAYASHI KIMITO PIONEER WORKS, MATSUMOTO ISAO PIONEER WORKS C, NAGASHITA TSUNEYOSHI PIONEER W, NAMIKI AKIO PIONEER WORKS CORP, NIINUMA SUSUMU PIONEER WORKS C, NOMURA ISAMU PIONEER WORKS COR, SHIMOKAWA KAZUTO PIONEER WORKS, WATANABE YOSHIHIKO PIONEER WOR, ABE, HIROYUKI PIONEER WORKS CORPORATION, AOYAGI, YOSHIO PIONEER WORKS CORPORATION, HAENO, AKIRA PIONEER WORKS CORPORATION, IIJIMA, TAKAYUKI PIONEER WORKS CORPORATION, KIMURA, TOSHIYUKI PIONEER WORKS CORPORATION, KOBAYASHI, KIMITO PIONEER WORKS CORPORATION, MATSUMOTO, ISAO PIONEER WORKS CORPORATION, NAGASHITA, TSUNEYOSHI PIONEER WORKS CORPORATION, NAMIKI, AKIO PIONEER WORKS CORPORATION, NIINUMA, SUSUMU PIONEER WORKS CORPORATION, NOMURA, ISAMU PIONEER WORKS CORPORATION, SHIMOKAWA, KAZUTO PIONEER WORKS CORPORATION, WATANABE, YOSHIHIKO PIONEER WORKS CORPORATION
Applicants: PIONEER ELECTRONIC CORP, PIONEER ELECTRONIC CORPORATION
Application year: 1986
Main classification: H02P7
IPC codes: H02P7