APPARATUS FOR OBTAINING DERIVATIVE OF CHARACTERISTIC CURVE, AND METHOD OF CONTROLLING OPERATION, OF ELECTRONIC DEVICES

Abstract: A novel technique for stabilizing an electronic device (40), e.g. for stabilizing the bias current of a semiconductor laser at or near the lasing threshold of the device, comprises deter­ mining a derivative (in 100) of a variable characteristic of the device operation, e.g. the voltage across the laser, with re­ spect to a parameter, e.g. the laser bias current. The derivative is determined by injecting one or more nonsinusoidal ac cur­ rent (I TEST ) components into the laser, and phase-sensitively detecting the voltage changes at the ac frequencies. An error voltage obtained (by 101-104) from the derivative is used to control the bias current (in 105). The derivative extraction method can also be embodied in measurement apparatus, and is useful in other process control applications.

Description: This invention relates to apparatus for obtaining a derivative of a characteristic curve, and to methods of controlling the operation, of electronic devices, e.g. semiconductor lasers. Semiconductor laser diodes are frequently used as radiation sources in optical fiber communications, as well as in other fields of technology. In many cases, the laser is operated in a pulse mode, e.g., having a two-level output, with the levels corresponding to logical one and logical zero, respectively. For instance, digital laser transmitters for optical communications frequently are driven by a high frequency modulation current superimposed on a dc bias current. The magnitude of the dc bias current is typically chosen so as to bias the laser near threshold. This practice results in a laser output such that the logic zero corresponds to no, or essentially no, radiation emitted from the laser, and the logic one level to a finite radiation intensity emitted from the laser. The dc bias current serves several purposes, but foremost among these are the functions of minimizing turn-on delay and minimising the amount of current that must be switched at the data rate. Because of possible drift in laser threshold due to aging and temperature variations, it is typically necessary to stabilize the bias current using feedback. Several methods are known for achieving stabilization. In systems applications, it has been the practice to use optical feedback, with an external photodetector either at the backside of the laser or at the front, using an optical fiber tap. See, for instance, pp. 182-186 of the article by P.W. Shumate and M. DiDomenico, "Lightwave Transmitters", of Semiconductor Devices for Optical Communications , H. Kressel, editor, second edition, Springer-Verlag, 1982. It is to be noted that such prior art stabilization schemes hold the light output of the laser constant, but do not necessarily stabilize the bias current at the desired level. This is a significant shortcoming, since several transmitter parameters, in addition to the extinction ratio, depend strongly on whether the laser is biased above, at, or below threshold. For example, when the laser is biased below threshold, extra turn-on delay typically occurs, leading to data pattern-dependent modulation errors in high speed systems. Also, in single-frequency laser transmitters, frequency shifting may occur as the modulation current drives the laser through threshold. And furthermore, in single loop stabilization systems, the efficiency of feedback in controlling laser variations is typically reduced when bias is sub-threshold. For a discussion of these points, see, for instance, G. Arnold and P. Russer, Applied Physics , Volume 14, November 1977, pp. 255-268; R.A. Linke, Electronics Letters , Volume 20, No. 11, May 1984, pp. 472-474; and R.G. Swartz and B.A. Wooley, Bell System Technical Journal, Volume 62, No. 7, Part 1, September 1983, pp. 1923-1936. On the other hand, when the laser is biased above threshold, reduced extinction ratio results. Also, when the laser is heated, or it ages, the threshold tends to increase so that a bias level initially above threshold can drop below it. Based on these and other considerations, it would be advantageous to have available a method for stabilizing of the bias point at or near threshold. The invention will be described by way of example with reference to the accompanying drawings, in which: FIGS. 1 to 3 show experimentally determined curves of first, second, and third derivatives, respectively, of the voltage across a semiconductor injection laser with respect to the forward current through the device, as a function of the forward current; FIGS. 4 to 6 show, schematically and in block diagram form, exemplary means for determining the first, second, and third derivative, respectively; FIGS. 7 to 9 schematically depict test signals that can advantageously be used in the determination of the second, third, and fourth derivative of the voltage as a function of the current, respectively; FIG. 10 shows, also schematically and in block diagram form, an exemplary means for controlling the bias current through a device; FIGS. 11 and 12 together depict on exemplary circuit for stabilizing the bias current of a semiconductor laser; FIG. 13 schematically depicts a communication system embodying the invention; and FIG. 14 schematically depicts a generalized process with a control system embodying the invention. Analogous features occurring in different figures are generally identified by the same reference numeral. As is well known, the dynamical resistance of a laser, i.e., the first derivative of its V/I characteristic, typically has a "kink" at threshold. This is illustrated in FIG. 1, which shows the quantity dV/dI as a function of the forward current through a semiconductor laser. Section 10 of the curve is the dynamical resistance below lasing threshold, and section 11 above losing threshold, with 12 referring to the "kink" in the dynamical resistance which coincides with the losing threshold. See, for instance, R.W. Dixon, Bell System Technical Journal, Volume 55, No. 7, September 1976, pp. 973-980. First and second derivative measurements of semiconductor laser I/V characteristics have been reported. These measurements were typically accomplished by injecting a sinusoidal low-frequency test current ΔI·cos(ωt) into the laser. The first derivative dV/dI was measured by synchronously detecting the component of the voltage response at frequency ω. The second derivative d²V/dI² was measured similarly by detecting the component of the voltage response at 2ω. Higher order derivatives can, in principle, be measured by detecting additional harmonics of the test response. Unfortunately, the unavoidable presence of harmonic distortion in the signal source leads to difficulty in accurately measuring second or higher order derivatives. Furthermore, this method requires relatively elaborate instrumentation. Methods are known for tracking and stabilizing the laser bias at threshold. For instance, D.W. Smith and P.G. Hodgkinson, Proceedings of the 13th Circuits and Systems International Symposium, Houston, 1980, pp. 926-930, describe an optical method that relies on synchronous optical detection of small laser test currents and which, for optimum performance, requires mixing of the low-frequency test signal with the high speed data. An all-electronic method of threshold detection was disclosed by A. Albanese, Bell System Technical Journal , Volume 57, No. 5, May-June 1978, pp. 1533-1544. This method relies on modeling the laser electrically as two series components, the first being a constant resistance, and the second having a diode-like characteristic, with a junction voltage saturating at laser threshold. Using an operational amplifier, the resistive component is balanced out. An error voltage signal is then generated by measuring the change in junction voltage in response to the data modulation current. When the junction is saturated, the modulation current should induce no change in junction voltage. With sub-threshold bias, however, a detectable change in junction voltage should be observable. The error signal produced by the modulation current can then be used in a feedback circuit to adjust the bias current to a level close to, but below threshold. The above method requires the detection of the modulation signal by the feedback circuit, necessitating the use of high frequency electronic circuitry. The method is also sensitive to the frequency of occurrence of logic ONE pulses in the data, requires careful tuning out of the laser series resistance, which must track with temperature, and is hampered especially by the fact that many lasers do not show strong junction voltage saturation at threshold. Canadian patent 1,101,923 discloses a method of operating an injection laser that involves applying a single component sinusoidal test signal of frequency f test and a bias current to the device, monitoring the level of the component of the light output of the laser at a frequency that is a harmonic of f test , and adjusting the bias current in accordance with the level of the component. U.S. patent 4,106,096 discloses a dual-frequency technique for electronically measuring a higher order derivative of the I/V characteristic of a semiconductor device. The technique involves applying a test signal to the device, the test signal having two phase-locked sinusoidal components of frequency f₁ and f₂, respectively. It further involves detecting the response of the device to the test signal at the difference frequency f₁-f₂ . As shown by the above discussion, such methods for stabilizing the bias current of a laser at or near threshold have considerable drawbacks. Due to the importance of reliably stabilizing the bias current of a laser at or near threshold, over a wide range of operating temperature and other operating conditions, a simple and inexpensively implemented method that is substantially free of the drawbacks associated with such methods would be of considerable interest. According to one aspect of this invention there is provided apparatus as claimed in claim 1. According to another aspect of this invention there is provided a method of controlling the operation of an electronic device, as claimed in claim 13. In its application to a laser the invention relies on the electrical characteristics of the laser, but does not require saturation of the laser junction voltage, and is insensitive to both laser series resistance and data pattern. In one embodiment of the invention, controlling an electronic device includes dithering the bias with the nonsinusoidal test signal and synchronously detecting the responsive signal. For instance, if the bias is a bias current, the test signal is advantageously a test current having first and second nonsinusoidal (preferably square wave) components, whereby the derivative of the V/I characteristic can be determined at two points close to the bias point, and d²V/dI² determined in a way that is analogous to the fundamental definition of the second derivative. Still higher derivatives can easily be obtained by extension of this procedure, including addition of further nonsinusoidal components to the test signal. The invention can also be embodied in apparatus for measuring a device characteristic, e.g., a higher order derivative of the voltage across an electronic device (for instance, a semiconductor laser or a tunneling diode) as a function of the current through the device, or of the light output of a semiconductor laser as a function of bias current. The invention may also be embodied in a communication system comprising a semiconductor laser radiation source, an optical fiber transmission medium, means for modulating the output of the laser at a signal frequency, means for coupling the output radiation into the optical fiber, and means for detecting the radiation, after transmission through a length of fiber. The laser has a V/I characteristic that relates the voltage across the laser to the current therethrough. The system also comprises means for providing a bias current to the laser, and means for changing the current through the laser. Furthermore, the system comprises electronic means for regulating the bias current, typically such that the bias current is maintained equal to the current corresponding to a particular point on the V/I characteristic, e.g., at or near the lasing threshold of the device. The current that corresponds to the laser threshold typically changes as a function of temperature and/or time. It is to be emphasized that in a system embodying the invention the bias current is not necessarily maintained constant, but instead, the system is regulated such that the bias current tracks the threshold current, or other appropriate point on the V/I characteristic. The means for regulating the bias current comprise first means for providing a test current to the laser, the test current comprising a nonsinusoidal first component of frequency f₁ substantially lower than the signal frequency, and also for obtaining a first signal that is substantially proportional to the test current-caused variation of the voltage across the laser. The regulating means further comprise second means for obtaining a second signal that is substantially proportional to a derivative of the V/I characteristic at the point corresponding to the bias current provided to the laser, third means for obtaining a third signal that is substantially proportional to the difference between the second signal and a reference signal, and fourth means for changing the bias current in response to the third signal. Controlling the operation of a communication system in this manner is a particular example of a broader process control technique. In general, a process can be described by specifying one or more variables X i (i = 1, 2, ...) that are functions of one or more parameters Y j (j = 1, 2, ...). Illustratively, a variable can be the rate of electrodeposition of a metal, and the parameters can be bath temperature, concentration of various ions in the bath, etc. The general technique comprises determining a derivative of a process variable as a function of a process parameter, i.e., determining d n X i /dY n / j(n = 2,3...). This is done in a manner akin to the method for determining d n V/dI n described herein, by varying the parameter Y j nonsinusoidally and measuring the resulting variations in X i . Typically, sensors that have an electrical output are available for measuring the value of any appropriate parameter and/or variable, and from the sensor outputs an electrical quantity proportional to d n X i /dY n / j is determined, e.g., by a method analogous to the derivative extraction method disclosed herein. This electrical quantity can then be compared with a target value, an error signal formed, and the error signal used to adjust one or more process parameters. Such derivative-based process control can be used independently, or in conjunction with known feedback control methods that do not involve derivative determination, and is advantageously applied in systems that exhibit a strong nonlinearity or even discontinuity in one of their process variables X i . Such nonlinearities typically are enhanced in higher order derivatives d n X i /dY n / j and may be advantageously used for process control. In stabilizing a semiconductor laser, for example, the invention may use a novel real time digital technique for determining voltage derivatives d n V/dI n (n = 2, 3, ...) that employs synchronous data detection and precisely timed phase reversals of a synchronizing reference clock. The novel technique has a theoretical accuracy that is nearly identical to the theoretically achievable accuracy of the conventional "harmonic distortion" method, and can be easily and inexpensively implemented with readily available, integrated circuitry. It can easily be used in, inter alia, optical communication systems that use semiconductor lasers as radiation sources. Unlike the known harmonic detection scheme, the present invention, to be referred to as "Phased Digital Derivative Extraction" (PDDE), uses test currents that need not be sinusoidal, and which preferably are square wave currents. This is a significant departure from the prior art, which results in the possibility of using well established digital technology in the implementation of the method. We will next demonstrate the principles of PDDE by reference to the determination of the first order voltage derivative, followed by discussion of the determination of second and third order voltage derivatives. From this, the general principle for determining any order derivative will emerge. The first voltage derivative versus current is measured by applying a small amplitude (typically square wave) test current ΔI₁ of frequency f₁ to the laser, while (typically) slowly sweeping the bias current I b . The voltage change ΔV₁ in response to the test current is measured synchronously. The first voltage derivative (which is the dynamic resistance R L ) is essentially equal to ΔV₁/ΔI₁. An exemplary circuit for determining ΔV₁ in response to the test current is schematically depicted In FIG. 4. Current source 41 causes dc bias current I b to flow through laser 40. Clock 42 generates a square wave clock signal of frequency f₁. The output of 42 is applied to current source 43 which causes test current ΔI₁ of frequency f₁ to be superimposed upon the bias current. The dc component of the voltage across 40 is blocked by means of high pass filter 44, and the ac component of the laser voltage is fed to the signal input of synchronous detector 45. The synchronous detector can be thought to comprise a pair of amplifiers A1 and A2, and a comparator C1. Synchronous detectors are well known in the art and are frequently referred to as double balanced mixers. The amplifiers have complementary gain characteristics, i.e., A1 has gain of +X and A2 has gain -X. The clock output is also connected to the non-inverting input of C1, with a reference voltage supplied to the reference input, such that the output of the comparator switches at frequency f₁, causing the output of the synchronous detector to be switched between A1 and A2 at the same frequency, depending on the polarity of the comparator output. The output of the synchronous detector is fed to low pass filter 46 to take the dc average of the output. The dc average is proportional to ΔV₁, and thus proportional to R L . When scaled by the proportionality factor ΔI₁ and the gain factor X/2, it is the voltage derivative of the laser. In a circuit as depicted In FIG. 4, only signal components varying at the same frequency as the test current add in phase at the output. All other components are mixed to some nonzero frequency and then eliminated by the low-pass filter. From the definition of the derivative, dR L /dI ∼ ΔR L /ΔI = (R L(A) -R L(B) )/ΔI₂, where subscripts A and B identify two points on the I/V characteristic of the device that differ by ΔI₂, and R L(A,B) = ΔV 1(A,B)/ ΔI₁, thus: d²V/dI² ∼ (ΔV 1(A) -ΔV 1(B) )/ΔI₁ΔI₂. The second voltage derivative can thus be determined by applying a two-component test current to the device. One of the components, of amplitude Δ+₁, has frequency f₁ and is employed as previously described. The other component has amplitude ΔI₂ and frequency f₂ and is employed to establish two measurement points A and B, separated by ΔI₂. As before, these components typically are nonsinusoidal, advantageously being square waves. For instance, the former might be a 1 kHz dither of amplitude 500 µA, while the latter could be a 125 Hz dither of amplitude 1 mA. FIG. 7 shows an exemplary test current for use in determination of the second voltage derivative. Superimposed on the bias current I b is a slow square wave of frequency f₂ and peak-to-peak amplitude ΔI₂, as well as a fast square wave of frequency f₁ and peak-to-peak amplitude ΔI₁. The "+1" and "-1" shown on the time axes of FIGS. 7, 8 and 9 indicate the cosine of the phase difference between the test current and the comparator input signal of the synchronous detector. An exemplary circuit for determining the second voltage derivative of a laser V/I characteristic is shown schematically in FIG. 5. The close similarity of this circuit to the previously described circuit for determining the first derivative is apparent. The normal output of clock 42 is fed to ÷ N digital counter 51, voltage scaling/summing junction 52, and the A-input of 2→1 multiplexer 53. The purpose of 51 us to generate the ΔI₂ signal at frequency f₂. Clock 42 also has inverted output 50 which is fed to the B-input of 53. The multiplexer steers either the normal or the inverted clock signal to the comparator input of synchronous detector 45. The selection of the normal or inverted clock signal is controlled by the signal at frequency f₂, shifting the phase of the comparator input by 180⁰ at intervals of 1/2f₂. This has the effect of multiplying the detector input by both an f₂ and an f₁ component, resulting in multiplication of any voltage component at frequency f₁ by +1 during the interval 0<t<1/2f₂, and by -1 during the interval 1/2f₂ <t< 1/f₂. The detector output, after lowpass filtering, is proportional to ΔV 1(A) -ΔV 1(B) , and the output, after normalization by the product ΔI₁ΔI₂, is proportional to the second voltage derivative. The third voltage derivative can be computed in similar fashion. Selecting four points A, B, C and D on the I/V curve where resistance is to be measured, with, for simplicity, points B and C chosen to coincide, it is easy to show that An exemplary test current useful for determining the third derivative is illustrated in FIG. 8. The waveform consists of four components: a dc bias I b , and three square wave components. One of the ac components has frequency f₂/2, another has frequency f₂, and the third has a frequency f₁, with f₁ > f₂. As illustrated, the former two components both have peak-to-peak amplitude ΔI₂, and the latter ΔI₁. Also, as illustrated, f₁ = 4f₂. It will be appreciated that these relationships are exemplary only. Indicated on FIG. 8 are also the above referred to four time intervals. FIG. 6 schematically illustrates an exemplary circuit capable of determining the third voltage derivative of the V/I characteristic of a device 40. The similarity of this circuit to the one illustrated in FIG. 5 is evident. The sole difference is a digital ÷ 2 counter following 51. The output of 60 changes once during a complete cycle in f₂. When the output of 60 is summed with the signals varying at f₂ and f₁, the staircase function of FIG. 8 is generated. The signal detection is synchronous as before, and the subtraction operation required for determining the third derivative will be accomplished by clocking the synchronous detector 180⁰ out of phase during the B and C intervals relative to the A and D intervals. From the preceding discussion, it is apparent that derivatives of arbitrary order can be produced. For example, a fourth order voltage derivative can be generated using the exemplary test current illustrated in FIG. 9, with the appropriate phasing of the detecting mixer. The current illustrated in FIG. 9 consists of four square waves superimposed on the dc bias current. The exemplary ac components have frequenices 3f₂, 2f₂. f₂, and f₁. It is also possible to measure several derivatives simultaneously, by using the same test current and dedicated mixers. For example, using the test current illustrated in FIG. 8 with test intervals, respectively, A, B, D and C, dV/dI is measured by applying +1+1+1+1 phasing to the mixer, d²V/dI² by applying -1-1+1+1 phasing, and d³V/dI³ by applying +1-1-1+1 phasing. This scheme also can be extended to derivatives of order higher than the third, as will be apparent to those skilled in the art. Mathematical analysis of the PDDE method shows that the method in principle can be highly accurate, with the magnitude of the error increasing as the order of the derivative increases, and as the ratio of ΔI₁ and/or ΔI₂ to I b increases. For instance, for an exponential I/V curve, when ΔI₁ is equal to 0.5 mA and ΔI₂ is equal to 1mA, the error can be less than 1% for I b greater than 2mA for the first derivative, 6mA for the second derivative, and 11mA for the third derivative. Although the PDDE method can be used to form the voltage derivative of any I/V device characteristic, and can, in principle, be used to determine the derivatives of one system parameter with respect to another system parameter, a particularly advantageous application of the method is in the stabilization of a semiconductor laser at or near the threshold of the laser, as will now be described. FIG. 2 shows d²V/dI² of a laser V/I characteristic, as a function of I b , with 21 corresponding to threshold. Points A and B on curve 20 identify bias currents near but slightly below and above threshold, respectively. A simple feedback scheme typically cannot determine the bias location relative to threshold (i.e. above or below), and therefore will not function properly for arbitrary initial bias. It is typically more advantageous to use a third derivative stabilization scheme, as can be seen from FIG. 3, which shows d³V/dI³ as a function of I b . Any of bias points F, G or I may be employed, with stable operation guaranteed for I b < I th for point F; for I b ∼ I th for point G; and for I b > I th for bias point I. The functional diagram of an exemplary feedback loop is shown in FIG. 10. Block 100 produces a voltage proportional to either the second or third order derivative, measured in response to a test current produced by current generator 43. The output voltage of 100 is inverted and summed with a reference voltage V R1 in 101, the resulting composite signal low-pass filtered in 102 to produce a dc derivative signal which is amplified in amplifier 103, summed with a second reference voltage V R2 in adder 104, and the voltage sum fed into a transconductance stage 105 to directly control the laser bias current I b . In a particularly advantageous stabilization scheme, the second derivative is used to locate threshold. This can be achieved as follows: when power is applied to the laser, the current is pre-biased to a point known to be above threshold, e.g. point C in FIG. 2. Using feedback with an error signal produced by the second derivative, the bias current is forced to converge to point B in FIG. 2, just above threshold. At this time, second derivative stabilization is complete, and laser bias could be maintained at this point. However, it may be desirable to go to third derivative stabilization and force the bias current to point G, precisely at threshold, in FIG. 3. In that case, one can switch to a third derivative-based error measurement, and apply a negative "kickback" current to push the laser bias into a region of third derivative stability near point G in FIG. 3. This kickback current will initially force the bias from somewhere near point I in FIG. 3 to somewhere between points J and H. Feedback will then cause bias to converge to point G. This advantageous combined second/third derivative stabilization scheme will now be described. With component 100 set to produce a voltage proportional to the second derivative, and power applied to the circuit of FIG. 10 at time t=0, I b = g m V R2 at t=0⁺, where g m is the transconductance of 105. Thus, V R2 is advantageously adjusted to bias the laser well above threshold at power-on, e.g. at point C of FIG. 2. For t much greater than the time constant of the feedback loop, I b ∼ I th + V R1 /(ΔI₁ΔI₂C₂) - K₂/C₂ where K₂ is the maximum negative excursion of the 2nd derivative peak at threshold, and C₂ is the positive slope of the derivative just above threshold, and all other quantities have their previously defined meaning. V R1 is advantageously chosen to bias the laser slightly above threshold, e.g. at point B in FIG. 2. If, at time t'=0, after attainment of a stable second derivative stabilization point, the operation of component 100 is switched to provide an output proportional to the third derivative, then at time t'=0⁺, This shows that I b at changeover depends on the second derivative bias point, so V R2 must be adjusted to substantially instantaneously move bias into the proper zone of third derivative convergence. This typically requires about 1mA negative current shift. When t'→∞ then generally where C₃ is a constant proportional to the slope of the third derivative curve at threshold. For stabilization at threshold, V R1 thus should be set equal to zero. An exemplary laser stabilization circuit was implemented, using only standard, commercially available, components. The ICs used are listed in Table I. INTEGRATED CIRCUIT CATALOG Component Description IC1 LM555 Timer IC2 4013 CMOS Dual D Flipflop IC3 4024 CMOS Counter IC4,IC4' 14538 CMOS Monostable Multivibrator IC5 4016 CMOS Quad Analog Switch IC6 AD63C Synchronous Modulator/Demodulator IC7 AD63C Synchronous Modulator/Demodulator IC8 LF347 Quad Op-Amp IC9 LF353 Dual Op-Amp IC10 LF351 Op-Amp IC11 LF351 Op-Amp IC12 LF353 Dual Op-Amp IC13 502BW Western Electric Building Block The portion of the circuit of FIG. 6 that is enclosed by dashed line 61 is shown in greater detail in FIG. 11. Circuit details that are well known to, or easily derived by, those skilled in the art are generally not shown. Such detail includes, for instance, pin numbers of ICs, supply voltages, some blocking or by-pass capacitors, some current limiting or voltage dividing resistors, and the like. The circuit of FIG. 11 can operate in two modes: second derivative and third derivative. The operating mode is determined by monostable IC4 and IC4' and is indicated by "STATUS OUT". When power is applied, IC4 generates a 30 second power-on pulse. During this time, STATUS is high, indicating second derivative operation. At the conclusion of this pulse, STATUS goes low, indicating third derivative operation. The circuit may be forced into second derivative operation by closing switch SW1. System timer IC1 operates at about 50 kHz. Flip-flop IC2A divides this clock down to 25 kHz with a 50% duty cycle. This is the critical timing component in the circuit, producing the clock signals f₁ and f₁ that yield ΔI₁ and synchronize the detector. The ÷ 4 output of counter IC3 produces the secondary clock f₂=f₁/4, approximately equal to 6.25 kHz. IC6 is a balanced modulator clocked by IC2B, with the frequency f₂ fed to its signal input. When the circuit is in a second derivative mode, flip-flop IC2B is held at 0, and the output of IC6 is the same as its input, i.e. a train of 0 V, + 15 V, 0 V, + 15 V, ....pulses at frequency f₁/4. When the circuit is in the third derivative mode, flip-flop IC2B is clocked by IC3 as a ÷ 2 counter (block 60 in FIG. 6). This causes the output of IC6 to invert every other pulse from IC3. In this case, the output of IC6 is the following repeating pattern: 0 V, + 15 V, 0 V, -15 V, 0 V, + 15 V, .... The part of the circuit enclosed by broken line 100 in FIG. 11 corresponds substantially to components 52 and 43 of FIG. 6. The negative input of operational amplifier IC10 is incrementally grounded by feedback around IC10. Test current ΔI₁ of peak-to-peak amplitude about 0.5 mA is produced by IC2A. Step current ΔI₂ is produced by IC6, its amplitude being about 1 mA. R₁₃ provides an offset to keep the total injected test current always positive. Finally, R₁₄ and the status signal balance the test current so that its average value is approximately 1 mA, independent of the operating mode. IC5 is a digital 2→1 multiplexer corresponding to 53 in FIG. 6, functionally equivalent to analog switches connected to form a single pole/double throw switch. The resulting "SYNCH" output is fed to the synchronous detector. Those skilled in the art will appreciate that an EXCLUSIVE OR gate could be substituted for IC5. FIG. 12 shows in greater detail the circuit corresponding substantially to the components enclosed by broken line 62 in FIG. 6, in addition to component 41 of that Figure. As will be appreciated by those skilled in the art, not all passive components required for satisfactory operation of the circuit are shown. The practitioner can easily provide such design detail. The core of this circuit section is the synchronous detector IC7, a precision component that provides the entire detector function on a single chip. V test is the voltage change across the laser in response to I test (see FIG. 11), and is advantageously measured as close as possible to the laser cathode. This voltage is amplified x5 by IC8A, high pass filtered and then amplified successively x20 and x3 by IC8B and IC8C, respectively. The output of IC8C is again high-passed (filter not shown) and then delivered to the signal input of IC7. IC7 is clocked by the SYNCH signal from IC5, with only the portion of V test at f₁ being in-phase with the synchronisation signal. Other frequency components are shifted with random phase to the output and are filtered out by a low-pass filter consisting of C8 and P30. IC9 adds offset V R1 to the output of IC7, and is adjustable by means of P₁ and P₂. Transistors Q₆, Q₇ and Q₈ form a circuit that allows both positive and negative offset adjustment. When the circuit is in second derivative mode, STATUS is high, and only P₁ controls the offset, and when the circuit is in third derivative mode, STATUS is low and only P₂ controls the offset. These offset controls typically must be adjusted properly for any given family of lasers or laser packages. JFET J₁ ground C₈ when power is off. The voltage on C₈ is directly proportional to -d²V/dI² or -d³V/dI³, depending on the circuit status. IC11 (which corresponds to amplifier 103 in FIG. 10) amplifies this voltage by x100. The output of IC11 is summed with two other signals at the input of IC12. The first of these signals is proportional V R2 , a dc voltage set by P₃ that determines the initial bias current in the laser when power is turned on. R₃₉ provides a "kickback" current of approximately 1 mA when the circuit switches from second to third derivative mode. This ensures that the laser bias is forced into the region of third derivative stability at the instant of second to third derivative changeover. Finally, IC12 supplies the actual bias current to the cathode of the laser. An alternative voltage-to-current converter that supplies current to the anode of the laser is also possible, as will be appreciated by those skilled in the art. The above described device stabilization technique has broad applicability, with laser stabilization being an exemplary application. A preferred application of a laser stabilized in this way is in a lightguide communication system. Such systems are well known and do not require detailed description. FIG. 13 schematically depicts such a system, in which laser 40 emits radiation 137 that is coupled by coupling means 130 into optical fiber 131 and is detected, after transmission through the fiber, by detector 132. A bias current is provided to the laser by means of bias supply 133, and a high frequency modulation current by means of modulation current source 132. Bias regulator 134 receives signals indicative of the laser voltage and of the dc bias by means of connections 135 and 136, respectively, and feeds a control signal to 133, whereby the dc bias level is controlled. As previously indicated, the instant technique for controlling the operation of a communication system is a particular example of a broadly applicable process control technique. FIG. 14 schematically illustrates this process control technique. Block 140 represents a general process, e.g. a chemical, electrochemical, pharmaceutical, or other technological process, which is characterised by variables X₁, X₂, ...., the variables being functions of parameters Y₁, Y₂, .... The value of a variable X i is sensed by means of sensor 141, and the value of a parameter Y j is sensed by sensor 142. The sensor outputs are supplied to block 145, which represents means for determining a derivative d n X i /dY n / j(n= 2, ...) from signals proportional to X i and Y j . The output of 145 is (typically) inverted by means of inverter 148 and summed with a reference signal in adder 149, and the resulting error signal 150 fed to "Y j -supply" 146, i.e. to means for providing an input to 140 that results in a particular value of parameter Y j . For instance, 146 could be means for changing the pressure in a reactor vessel. The output of 146 contains both a constant or slowly-varying value (the "bias") and nonsinusoidally varying components (the "dither"). The latter produce corresponding changes in X i , which are sensed by 141, and the dither-component of Y j is sensed by 142. An exemplary application of the control process is in vehicular control, e.g. in anti-lock braking systems for automobiles, trucks, trains, and the like. Other applications will be apparent to those skilled in the art. It is frequently desirable to measure a derivative, especially a higher order derivative, of a device characteristic, and PDDE can also be advantageously used for this purpose. The elements of the technique as recited, for instance, in the above description, except the means for obtaining an error signal and the means for changing the bias in response to the error signal, are also present in measuring apparatus embodying the invention. In addition, the measuring apparatus comprises means for displaying the value of the derivative, or other means for indicating the measurement result. Such means may comprise a visual read-out, electrical read-out means that provide a digital or analog signal, or go/no go indicators such as signal ights. Such means are conventional and well known in the art.

Claims: Apparatus for obtaining a derivative of a characteristic curve of an electronic device (40), comprising first means (41) for providing an electrical bias to the device to determine an operating point on the characteristic curve, second means (42,43,44) for providing a test signal to the device, and for obtaining a first signal responsive to the test signal provided to the device, the test signal comprising a component of frequency f₁ and a component of frequency f₂≠f₁, and third means (45,46) for obtaining from the first signal a second signal that is substantially proportional to a derivative of the characteristic curve at the operating point determined by the bias, CHARACTERIZED IN THAT the derivative is a Nth-order derivative, where N≧2, the test signal comprises a linear sum of N phase-locked components, any given component being a non-sinusoidal signal of frequency that differs from that of all other components of the test signal, the third means comprise multiplier means (45) for obtaining a third signal that is substantially proportional to the product of the first signal and a signal that is a product of said N components of the test signal, the output of said multiplier means being provided to low pass filter means (46), the output of said low pass filter means being the second signal, and the apparatus comprises utilization means responsive to the second signal. Apparatus as claimed in claim 1 wherein any given component of the test signal is substantially a square wave. Apparatus as claimed in claim 1 or 2 wherein the device is a semiconductor laser, and the characteristic curve is the V/I characteristic or the L/I characteristic, where V is the voltage across the laser, L is the light output of the laser, and I is the current through the laser, or the device is a tunnel diode. Apparatus as claimed in claim 1 or 2 wherein the characteristic curve is a V/I characteristic relating device voltage to device current, and the utilization means comprises means (101-104) for providing an error signal whose amplitude is related to the amplitude of the second signal, and means (105) for adjusting the electrical bias in response to the error signal. Apparatus as claimed in claim 4, wherein the device is a semiconductor laser, and the V/I characteristic relates the voltage across the laser to the current through the laser. Apparatus as claimed in claim 5 wherein the operating point is a point at which at least one derivative is nonzero. Apparatus as claimed in claim 6 wherein the operating point is at, or close to, the lasing threshold. Apparatus as claimed in claim 6 or 7 wherein the second signal is proportional to the third derivative of the characteristic during at least a part of the time the bias current is being adjusted. Apparatus as claimed in claim 8 wherein the second signal is proportional to the second derivative of the characteristic during a further part of the time the bias current is being adjusted. Apparatus as claimed in claim 6 or 7 wherein the second signal is proportional to the second derivative of the characteristic during at least a part of the time the bias current is being regulated. Apparatus as claimed in any preceding claim wherein the test signal has a third nonsinusoidal component of frequency f₃, with f₁ > f₂ > f₃. Apparatus as claimed in claim 11 wherein each said component of the test signal is substantially a square wave. A method of controlling the operation of an electronic device (40) having a V/I characteristic relating a device voltage to a device current, comprising a) providing, by first means (41), an electrical bias to the device to determine an operating point on the V/I characteristic, b) providing, by second means (42,43,44), a test signal to the device and obtaining a first signal responsive to the test signal provided to the device, the test signal comprising a component of frequency f₁, c) obtaining, by third means (45,46). from the first signal a second signal that is substantially proportional to a derivative of the V/I characteristic at the operating point determined by the bias, d) obtaining, by fourth means (101-104), an error signal related to the amplitude difference between the second signal and a reference signal, and e) adjusting the bias in response to the error signal, CHARACTERIZED IN THAT f) the derivative is a N-th order derivative, where N≧2, g) the test signal comprises a linear sum of N phase-locked components, any given component being a non-sinusoidal signal of frequency that differs from that of all other components of the test signal, and c) comprises obtaining, by means that comprise multiplier means (45), a third signal that is substantially proportional to the product of the first signal and a signal that is a product of said N components of the test signal, and providing the third signal to low pass filter means (46), the output of said low pass filter means being the second signal. A method as claimed in claim 13 wherein any given component of the test signal is substantially a square wave. A method as claimed in claim 13 or 14 wherein the second signal is proportional to the second derivative of the characteristic during at least a part of the time the operation of the device is controlled. A method as claimed in claim 13 or 14, wherein the second signal is proportional to the third derivative of the characteristic during a part of the time the operation of the device is controlled. A method as claimed in any one of claims 13 to 16 wherein the device is operated at a signal frequency that is substantially higher than f₁. A method as claimed in claim 17 wherein the device is a semiconductor laser, the bias is a bias current, and the bias current is modulated at the signal frequency.

Inventors: SWARTZ, ROBERT GERALD
Applicants: AMERICAN TELEPHONE AND TELEGRAPH COMPANY
Application year: 1986
Main classification: H04B10
IPC codes: H01L21, H01L29, H04B10, H01S5