CONTROL METHOD FOR CYCLOCONVERTER AND CONTROL APPARATUS THEREFOR

Abstract: There is disclosed a control system for a cycloconverter of the circulating current type to produce an ac current of a variable frequency and a system therefore. This cycloconver­ ter (CC) is configured to connect a power factor compensat­ ing capacitor (CAP) at its receiving-end, thus to control a lagging reactive power due to a circulating current flowing in the cycloconverter (CC) so as to cancel a leading reactive power due to the power factor compensating capacitor (CAP). Such a control is carried out under the condition of an operation below the rated load. In contrast, under the condition of an operation above the rated load, a control is effected so that the circulating current flowing in the cycloconverted (CC) is substantially kept constant.

Description: The present invention relates to a cycloconverter system of the circulating current type to deliver a variable frequency ac current to a single- or polyphase load. A cycloconverter is an apparatus to directly convert an ac power having a constant frequency into an ac power having a different frequency. The drawback with the cycloconverter is that it is necessary to commutate thyristors serving as elements constituting the cycloconverter by a power supply voltage, with the result that a large reactive power is produced from a power source. Further, the reactive power always varies in synchronism with a frequency on the side of a load. As a result, this not only requires the capacitance of a power system equipment to be increased, but also gives various bad influences on electric equipment connected to the same system. To overcome this, a measure has been taken to install a reactive power compensating device at a receiving-end of a cycloconverter. However, such a measure results in large equipment, an increased area therefor and a high cost. In view of this, a reactive power compensative cycloconverter as disclosed in the Japanese Publication Tokkaisho No. 56-44382 etc. is proposed to solve the above-mentioned problem. Namely, a cycloconverter of the circulating current type is used to connect a power factor compensating capacitor at a receiving-end of the cycloconverter, thus to control a circulating current of the cycloconverter so that a leading reactive power by the power factor compensating capacitor and a lagging reactive power by the cycloconverter are cancelled with each other. The role of a reactive power compensating device which has been needed in the prior art is incorporated within the cycloconverter itself. As a result, the conventional reactive power compensating device is not needed. According to this, the cycloconverter can be of a small size and lightened, and its cost can be reduced. In the above-mentioned reactive power compensative cycloconverter system, the capacitance of a power factor compensating capacitor at the receiving-end is determined as a standard when the cycloconverter effects a rated operation. When an overload operation is expected, it is necessary to provide in advance a power factor compensating capacitors having a capacitance to cope therewith. In other words, if one attempts to affect a control such that an input power factor at a receiving-end is always maintained at 1, an output capacity of the cycloconverter is determined by the capacitance of the power factor compensating capacitor, with the result that an overload running which requires a capacity above the capacitance of the load capacitor is impossible. Further, when a power factor compensating capacitor having an excessive capacitance is connected in advance in anticipation of an overload operation, a circulating current which is to flow in a cycloconverter at the time of a rated load or a light load is increased. As a result, this leads to an increase in capacity of a converter or a power transformer, or will result in a system having a power efficiency due to an increase in loss. In US 4418380 a method and apparatus for controlling the circulating current of a cycloconverter is disclosed wherein the circulating currents of the cycloconverter in respective phases are distributed in such a way that the circulating current is small when the absolute value of the load current is large and vice versa. However, the maximum power outputted from the cycloconverter is limited by the capacity of the phase advancing capacitor and no provision is described for exceeding this limit. Thus, there are no provisions in the event the cycloconverter is caused to operate under overload conditions. An object of the present invention is to provide a control method for a cycloconverter of the circulating current type and a control apparatus therefor which makes it possible to effect a stabilized overload operation without increasing the capacitance of a power factor compensating capacitor. In accordance with the present invention, by making use of the advantage (high upper limit of an output frequency) of a cycloconverter of the circulating current type, an overload operation is carried out wherein a reactive power control is affected so that an input power factor is maintained at 1 until reaching a rated load. Accordingly, it is sufficient to provide for a power factor compensating capacitor having a capacitance enough to cancel a lagging reactive power given by the cycloconverter at the time of the rated operation. At the time of an overload operation, the reactive power control at a receiving-end stops control such that a minimum circulating current continues to flow in the cycloconverter. Thus, there is no possibility that a break or interruption of a circulating current occurs, thus enabling a stabilized overload operation. The present invention provides therefore a cycloconverter apparatus comprising: a cycloconverter of the circulating current type for connection to a single or poly-phase load, said cycloconverter being adapted to receive power from an AC power supply; a power factor compensating capacitor connected to a receiving end of said cycloconverter; means for determining a reactive power of the cycloconverter apparatus; a command value generator connected to the means for determining the reactive power and responsive to the reactive power determined for generating a command value; comparator means connected to the command value generator for comparing the circulating current flowing in the cycloconverter with the command value generated; and controlling means connected to the comparator means for controlling the circulating current flowing in said cycloconverter in response to the result of the comparison by the comparator means characterised by further comprising: command value adjustment means connected to the command value generator for changing the command value generated by said command value generator when an output current of said cycloconverter is greater than a rated value such that the command value is set and remains substantially constant regardless of variations in output current greater than the rated value whereby the circulating current flowing in said cycloconverter remains substantially constant. In a further aspect the present invention provides a control method for a cycloconverter of the circulating current type comprising the steps of: generating a circulating current command value in response to a reactive power at a receiving end of the cycloconverter; comparing a circulating current with a circulating current command value; and controlling the circulating current in response to the result of the comparison whereby a lagging reactive power of the cycloconverter and a leading reactive power of the power factor compensating capacitor connected to a receiving end of the cycloconverter are cancelled with each other when an output current of the cycloconverter is less than a rated value and characterised in that, when the output current is greater than the rated value, the circulating current command value is set and remains substantially constant by operation of command value adjustment means regardless of variation in the output current greater than the rated value whereby the circulating current remains substantially constant. In the drawings, Fig. 1 is a block diagram illustrating an embodiment of a cycloconverter system according to the present invention, Fig. 2 is a view showing the relationship between voltage and current vectors for explaining the operation of the system shown in Fig. 1, Fig 3 is a view showing the relationship of a load current I L versus a circulating current I O and a power factor at a receiving-end for explaining the operation of the system shown in Fig. 1, Fig. 4 is a view showing an output voltage wave-form of a cycloconverter of the circulating current type, Fig. 5 is a block diagram illustrating another embodiment of a cycloconverter system according to the present invention, Fig. 6 is a circuit diagram illustrating an example of a reactive power command value circuit provided in the system shown in Fig. 5, and Fig. 7 is a view showing the relationship of a load current amplitude I m versus a reactive power command value Q s * and a circulating current I o for explaining the operation of the system shown in Fig. 5. The present invention will be described in detail in connection with preferred embodiments with reference to attached drawings. Referring to Fig. 1, there is shown, in a block form, a first preferred embodiment of a cycloconverter system according to the present invention. The cycloconverter system including a power supply line BUS of a three phase power supply, a power factor compensating capacitor assembly CAP comprising delta or star connected capacitors, a power transformer TR, a cycloconverter CC of the circulating current type, and a load LOAD. The cycloconverter CC comprises a positive group converter SSP, a negative group converter SSN, and dc reactors L₀₁ and L₀₂ with intermediate taps. A control circuit for the cycloconverter CC comprises a load current detector CTL, an output current detector CT P of the positive group converter, an output current detector CT N of the negative group converter, a transformer PT S for detecting a three phase ac voltage at a receiving-end, a current transformer CT S for detecting a three-phase ac current, a reactive power computing circuit VAR, comparators C₁, C₂ and C₃, adders A₁, A₂ and A₃, control compensating circuits H Q (S), G O (S) and G L (S), a limiter circuit LIM, a circulating current setter VR O , an operational amplifier K₁ and phase control circuits PHP and PHN. Initially, the operation of a load current control will be described. A load current command I L * and a value detected of a current I L flowing in the load are input to the comparator C₃ to obtain an error ε₃=I L *-I L . The error ε₃ is input to the load current control compensating circuit G L (S) to carry out proportional amplification or integral amplification. In this example, the load current control compensating circuit G L (S) includes only a proportional element (gain K L ) for brevity of explanation. It is now assumed that an output signal from the circulating current control circuit G O (S) is sufficiently small. When the output signal is negligible, an input signal of the phase control circuit PHP for the positive group converter SSP is expressed as V cP =K L ·ε₃ and its output voltage is expressed as follows: V P = K V ·V S ·cos αP ∝ V αP . At this time, to the phase control circuit PHN for the negative group converter SSN, a sign V αN =K L ·ε₃ is obtained by passing an output signal K L ·ε₃ from the control compensating circuit G L (S) through the inverting operational amplifier K₁. Accordingly, an output voltage V N of the negative group converter SSN is expressed as V N = -K V ·V S ·cosα N ∝ -Vα N = V P , where K V denotes a conversion constant, V S a power supply voltage, and α P and α N firing phase angles. Namely, an ordinary operation is carried out with the output voltage V P of the positive group converter SSP and the output voltage V N of the negative group converter SSN being balanced with each other at the load terminal. In this instance, the relationship of α N = 180°-α P holds in connection with the firing phase angles of the two converters. To the load LOAD, V L = (V P +V N )/2 which is an average value of output voltages of the two converters is applied. When I L * > I L , the error ε₃ becomes a positive value to increase the output voltages V P and V N in a direction indicated by the arrow in the figure. Accordingly, the load terminal voltage V L is increased to increase the load current I L . In contrast, when I L * < I L , the error ε₃ becomes a negative value to produce the output voltages V P and V N in a direction opposite to the arrow in the figure. As a result, the load terminal voltage V L becomes a negative value to decrease the load current I L . For this reason, the relationship of I L *≒I L will be eventually established, thus placing the control system in a balanced condition. Further, when an operation is effected to sinusoidally change the current command I L *, the error ε₃ varies according to this. Accordingly, the above-mentioned firing phase angles α P and α N are controlled in a manner that a sinewave current I L flows in the load. In such an ordinary operation, the voltages of the positive group converter SSP and the negative group converter V N are balanced with each other, with the result that little circulating current I O flows. Subsequently, the control operation of the circulating current I O will be described. The circulating current I O of the cycloconverter is detected as follows. The circulating current I O is determined by obtaining the sum of a value detected of the output current I P of the positive group converter SSP to a value detected of the output current I N of the negative converter SSN, subtracting an absolute value of a value detected of the toad current I L from the sum, and multiplying it by one-half. The relationship is expressed as follows: I O = (I P +I N -|I L |)/2 The circulating current I O thus obtained is compared with the command value I O *. An error ε₂=I O *-I O is input to the next stage of the circulating current control compensating circuit G O (S). The error ε₂ undergoes proportional amplification or integral amplification. In this example, it is assumed for brevity of explanation that G O (S)=k O , i.e., the circulating current control compensating circuit G O (S) includes only a proportional element. Thus, an output signal of the circulating current control compensating circuit G O (S) is input to the adders A₂ and A₃. Accordingly, input voltages V P and V N to the phase control circuits PHP and PHN are respectively expressed as follows: V P = K L ·ε₃ + K O · ε₂ , and V N = -K L ·ε₃ + K O ·ε₂. Accordingly, the relationship of α N ≒180°-α P cannot be maintained. The output voltage V P of the positive group converter SSP and the output voltage V N of the negative group converter SSN are unbalanced by a value proportional to K O ·ε₂. Its difference voltage (V P -V N ) is applied to the dc reactors L₀₁ and L₀₂, with the result that the circulating current I O flows. When I O * > I O , the error ε₂ becomes a positive value. This increases the output voltage V P and decreases the output voltage V N . Accordingly, the difference voltage (V P -V N ) becomes a positive value to increase the circulating current I O . In contrast, when I O * < I O , the error ε₂ becomes a negative value. This decreases the output voltage V P and increases the output voltage V N . Accordingly, the difference voltage (V P -V N ) becomes a negative value to decrease the circulating current I O . The relationship of I O *≒I O will be eventually established, thus placing the control system in a balanced condition. In such a circulating current control, the output voltages V P and V N of the positive and negative group converters vary. However, since the load terminal voltage V L is obtained as an average value of the output voltages V P and V N , such a change does not affect the load current control. On the other hand, the reactive power control is carried out as follows. At the receiving-end, the three phase current detector CT S and the three phase voltage detector PT S are provided. The reaction power Q is computed by the reactive power computation circuit VAR. The command value Q* for the reactive power control is ordinarily set at zero. The comparator C₁ produces an error ε₁=Q*-Q . The reactive power control compensating circuit H Q (S) ordinarily employs integral element in order to allow the steady-state error ε₁ to be zero. By passing an output from the control compensating circuit H Q (S) through the adder A₁, the above-mentioned circulating current command value I O * is obtained. Fig. 2 is a view showing the relationship between voltage and current vectors at the receiving-end wherein the vector relationship represents a single-phase of the three phase power supply. In this figure, V S represents a power supply voltage, I cap a leading current flowing in the power factor compensating capacitor, I CC an input current of the cycloconverter, I S an effective value of the input value, and I REACT a reactive value of the I S . Further, I SSP and I SSN represent input currents of the positive and negative group converters, respectively. Where a current conversion factor of the converter is denoted by K₁, the above-mentioned input currents I SSP and I SSN are expressed as follows: I SSP = K₁·I P , and I SSN = K₁·I N . This vector view indicates the condition where the load current I L is delivered from the positive group converter SSP. The output current I P of the positive group converter is expressed as I P =I L +I O and the output current I N of the negative group converter is expressed as I N =I O . The input current I CC of the cycloconverter is obtained as the vector sum of the input currents I CC and I SSP . The effective value I S and the reactive value I REACT of the input current I CC can be expressed as follows: where the relationship of α N ≒180-α P is introduced in the above deviation. Namely, the circulating current I O flowing in the cycloconverter does not affect the effective value I S on the input side, but affects the reactive value I REACT . By effecting a control of the circulating current I O so that the concerned reactive current I REACT is always equal to the leading current I CAP of the power factor compensating capacitor, the current I S delivered from the power supply always becomes effective component, thus maintaining the fundamental wave power factor at 1. Turning to Fig. 1, when Q* > Q (the lag becomes positive), the error ε₁=Q*-Q becomes positive to increase the circulating current command value I O * through the control compensating circuit H Q (S). Accordingly, the actual value I O ≒I O * of the circulating current is increased, thus to increase the lagging reactive current value I REACT of the input current I CC flowing in the cycloconverter. As a result, the lagging reactive power Q at the power receiving-end is increased, effecting a control so that Q≒Q* is attained. In contrast, when Q* < Q, the circulating current I O is decreased, with the result that the delayed reactive power Q is also decreased. Thus, the cycloconverter is placed in a balanced condition where Q≒Q*. Fig. 3 represents the relationship of the load current I L versus the circulating current I O and the power factor P.F. When the load current I L is zero, only the circulating current I O flows in the cycloconverter CC to maintain the relationship I cap =I REACT . Accordingly, the value of circulating current becomes large. As the value of the load current I L increases, the value of the circulating current I O is decreased, thus maintaining the input power factor at 1. When the load current exceed a rated current by further increasing the load current I L , the leading reactive power given by the power factor compensating capacitor CAP lacks with respect to the lagging reactive power given by the cycloconverter, resulting in the lagging power factor at the receiving-end. Accordingly, the value of the error ε₁=Q*-Q in the reactive power control becomes negative and the output of the control compensating circuit H Q (S) also becomes negative. However, when the output of H Q (S) represents a negative value by the limiter circuit LIM, the output I* QQ of the limiter circuit LIM becomes zero. On the other hand, another circulating command value ΔI O * is given by a circulating current setter VR O . The current command value ΔI O * is set at a value such that circulating current of the cycloconverter CC is not interrupted. Accordingly, the output I O *=I* QQ +ΔI O * of the adder is actually given as the circulating current command value. An actual circulating current I O is controlled in accordance with the output of the adder. Accordingly, even when the load current I L increases to exceed a rating value, a control is effected so that the circulating current I O of the cycloconverter CC is not interrupted. In such a case, the power factor at the receiving-end is not equal to 1. Accordingly, according as the load current I L increases, there occurs a condition where the power factor lags. This is no problem in the system where the overload running is seldom carried out. Fig. 4 shows an output voltage waveform of a cycloconverter of the circulating current type having twelve control phases when the setting is made such that an output frequency f₀ is 1.2 times larger than an input frequency f₁. In this figure, V i (thin solid line) represents an input voltage, V P (thick single dotted lines) an output voltage of the positive group converter, V N (thick double dotted lines) an output voltage of the negative group converter, V R (thick dotted lines) an output voltage required, and V O (thick solid line) an output voltage (which is applied to the load) actually produced by the cycloconverter. The output voltage V O is obtained as an average of the output voltages V P and V N and represents a stair-stepped waveform. The output voltages V P and V N are created by connecting portions (segments) of the input voltage Vi to each other, respectively. When attention is drawn to only each voltage, an uncontrollable period occurs in the same manner as the cycloconverter of a non-circulating current type does. However, when the output voltage V P is in an uncontrollable condition, the output voltage V N functions so as to compensate for the lack of the ability of the former. On the other hand, in the range where the output voltage V N is in an uncontrollable condition, the output voltage V P functions so as to compensate for the lack of the ability of the former. As a result, the output voltage V O can be controlled so that it follows the command value V R . Namely, the normal operating mode of the cyclo-converter of the circulating current type is to operate both the positive group converter SSP and the negative converter SSN at the same time. Accordingly, if an operating mode is employed to activate only one of them, viz., to allow the circulating current to be zero, the cycloconverter has a sawtooth output waveform due to the operation of either one of the positive and negative group converters, with the result that many harmonics are included. Accordingly, the cycloconverter of the circulating current type employs an operating mode such that the circulating current simply flows therein. Fig. 5 is a block diagram illustrating another embodiment of a cycloconverter system according to the present invention. The cycloconverter system in this embodiment includes a power supply line BUS of a three phase power supply, a power factor compensating capacitor assembly CAP comprising delta or star connected capacitors, a power transformer TR, a cycloconverter CC of the circulating current type, and a load LOAD. The cycloconverter CC comprises a positive group converter SSP, a negative group converter SSN, and dc reactors L₀₁ and L₀₂ with intermediate taps. A control circuit for the cycloconverter CC comprises a load current detector CT L , an output current detector CT P of the positive group converter, an output current detector CT N of the negative group converter, a transformer PT S for detecting a three phase ac voltage at a receiving-end, a current transformer CT S for detecting a three phase ac current, a reactive power computation circuit VAR, comparators C₁, C₂, C₃ and C₄, adders A₁ and A₂, a multiplier ML, control compensating circuits G N (S), H Q (S), G O (S) and G L (S), an absolute value circuit ABS, a reactive power command value circuit FQ, an operational amplifier K₁, and phase control circuits PHP and PHN. Here, reference is made to the case where a cycloconverter is applied to a speed control system for an electric motor. The output I m of the control compensating circuit G N (S) produced when a difference ε₄ between the speed reference N* and the actual rotational speed N is input to the control compensating circuit G N (S) corresponds to an amplitude of the motor current. The output I m is fed to the amplifier ML and is then multiplied by a sinewave signal sinωt synchronous with a speed electromotive force of the motor thereat. Thus, the load current command I L * is created as an output of the multiplier ML. Initially, the operation of a load current control will be described. A load current command I L * and a value detected of a current I L flowing in the load are input to the comparator C₃ to obtain an error ε₃=I L *-I L . The error ε₃ is input to the load current control compensating circuit G L (S) to carry out proportional amplification or integral amplification. In this example, the load current control compensating circuit G L (S) includes only a proportional element (gain K L ) for brevity of explanation. It is now assumed that an output signal from the circulating current control circuit G O (S) is sufficiently small. When the output signal is negligible, an input signal of the phase control circuit PHP for the positive group converter SSP is expressed as Vα P =K L ·ε₃ and its output voltage is expressed as follows: V P = K V ·V S ·cosα P ∝ Vα P . At this time, to the phase control circuit PHN for the negative group converter SSN, a signal Vα N =-K L ·ε₃ obtained by passing an output signal K L ·ε₃ from the control compensating circuit G L (S) through the inverting operational amplifier K₁ is input. Accordingly, an output voltage of the negative group converter SSN is expressed as V N = -K V ·V S ·cosα N ∝ -Vα N = V P , where K V denotes a conversion constant, V S a power supply voltage, and α P and α N firing phase angles. Namely, an ordinary running is carried out with the output voltage V P of the positive group converter SSP and the output voltage V N of the negative group converter SSN being balanced with each other at the load terminal. In this instance, the relationship of α N =180°-α P holds in connection with the firing phase angles of the two converters. To the load LOAD, V L =(V P +V N )/2 which is an average value of output voltages of the two converters is applied. When I L * > I L , the error ε₃ becomes a positive value to increase the output voltages V P and V N in a direction indicated by the arrow in the figure. Accordingly, the load terminal voltage V L is increased to increase the load current I L . In contrast, when I L * < I L , the error ε₃ becomes a negative value to produce the output voltages V P and V N in a direction opposite to the arrow in the figure. As a result, the load terminal voltage V L becomes a negative value to decrease the load current I L . For this reason, the relationship of I L *≒I L will be eventually established, thus placing the control system in a balanced condition. Further, when an operation is effected to sinusoidally change the current command I L *, the error ε₃ varies according to this. Accordingly, the abovementioned firing phase angles α P and α N are controlled in a manner that a sinewave current I L flows in the load. In such an ordinary operation, the voltages of the positive group converter SSP and the negative group converter V N are balanced with each other, with the result that little circulating current I O flows. Subsequently, the control operation of the circulating current I O will be described. The circulating current I O of the cycloconverter is detected as follows. The circulating current I O is determined by obtaining the sum of a value detected of the output current I P of the positive group converter SSP to a value detected of the output current I N of the negative converter SSN, subtracting an absolute value of a value detected of the load current I L from the sum, and multiplying it by one-half. The relationship is expressed as follows: I O = (I P +I N -|I L |)/2 The circulating current I O thus obtained is compared with the command value I O *. An error ε₂=I O *-I O is input to the next stage of the circulating current control compensating circuit G O (S). The error ε₂ undergoes proportional amplification or integral amplification. In this example, it is assumed for brevity of explanation that G O (S)=K O ·ε₂ i.e., the circulating current control compensating circuit G O (S) includes only a proportional element. Thus, an output signal of the circulating current control compensating circuit G O (S)=K O is input to the adders A₁ and A₂. Accordingly, input voltages V P and V N to the phase control circuits PHP and PHN are respectively expressed as follows: V P = K L ·ε₃ + K O ·ε₂, and V N = -K L ·ε₃ + K O ·ε₂. Accordingly, the relationship of α N ≒180°-α P can not be maintained. The output voltage V P of the positive group converter SSP and the output voltage V N of the negative group converter SSN are unbalanced by a value proportional to K O .ε₂. Its difference voltage (V P -V N ) is applied to the dc reactors L₀ ₁ and L₀₂, with the result that the circulating current I O flows. When I O * > I O , the error ε₂ becomes a positive value. This increases the output voltage V P and decreases the output voltage V N . Accordingly, the difference voltage (V P -V N ) becomes a positive value to increase the circulating current I O . In contrast, when I O * < I O , the error ε₂ becomes a negative value. This decreases the output voltage V P and increases the output voltage V N . Accordingly, the difference voltage (V P -V N ) becomes a negative value to decrease the circulating current I O . The relationship of I O *≒I O will be eventually established, thus placing the control system in a balanced condition. In such a circulating current control, the output voltages V P and V N of the positive and negative group converters vary. However, since the load terminal voltage V L is obtained as an average value of the output voltages V P and V N , such a change does not affect the load current control. On the other hand, the reactive power control is carried out as follows. At the receiving-end, the three phase current detector CT S and the three phase voltage detector PT S are provided. The reactive power Q S is computed by the reactive power computation circuit VAR. The command value Q* of the reactive power is ordinarily set at zero. The comparator C₁ produces an error ε₁=Q S *-Q S . The reactive power control compensating circuit H O (S) ordinarily employs integral element in order to allow the steady-state error ε₁ to be zero. The output of the reactive power compensating control circuit H Q (S) is the above-mentioned circulating current command value I O *. Fig. 2 is a view showing the relationship between voltage and current vectors at the receiving-end wherein the vector relationship represents a single phase of the three phase power supply. In this figure, Icap represents a leading current flowing in the power factor compensating capacitor, I CC an input current of the cycloconverter, I S an effective component of the input value, and I REACT a reactive component of the I REACT . Further, I SSP and I SSN represent input currents of the positive and negative group converters, respectively. Where a current conversion factor of the converter is denoted by K₁, the above-mentioned input currents I SSP and I SSN are expressed as follows: I SSP = K₁·I P , and I SSN = K₁·I N . This vector view indicates the condition where the load current I L is delivered from the positive group converter SSP. The output current I P of the positive group converter is expressed as I P =I L +I O and the output current I N of the negative group converter is expressed as I N =I O . The input current I CC of the cycloconverter is obtained as the vector sum of the input currents I CC and I SSP . The effective component I S and the reactive component I REACT of the input current I CC can be expressed as follows: where the relationship of α N ≒180-α P is introduced in the above derivation. Namely, the circulating current I O flowing in the cycloconverter does not affect the effective component I L on the input side, but affects the reactive component I REACT . By effecting a control of the circulating current I O so that the concerned reactive current I REACT is always equal to the leading current I CAP of the phase advance capacitor and the current I S delivered from the power supply always becomes effective component, thus maintaining the fundamental wave power factor at 1. Turning to Fig. 5, when Q S * > Q S (the lag is a positive value), the error ε₁=Q S *-Q S becomes positive to increase the circulating current command value I O * through the control compensating circuit H O (S). Accordingly, the actual value I O ≒I O * of the circulating current is increased, thus to increase the delay reactive current component I REACT of the input current I CC flowing in the cycloconverter. As a result, the delay reactive power Q at the power receiving-end is increased, effecting a control so that Q≒Q* is attained. In contrast; when Q* < Q, the circulating current I O is decreased, with the result that the delayed reactive power Q is also decreased. Thus, the cycloconverter is placed in a balanced condition where Q S ≒Q S * is attained. Then, assuming that an electric motor is employed as the load, the operation of the rotational speed control for the motor will be described. Where a dc motor is employed as the load, it can be said that the load LOAD in Fig. 1 corresponds to an armature winding of the dc motor. Further, where an ac motor is employed as the load, it can be said that the load LOAD in Fig. 1 corresponds to one phase of an armature winding of the ac motor. In this example, an explanation will be made in connection with the ac motor (synchronous motor). A rotational speed N of the motor is detected by a tachogenerator or a rotatable pulse generator etc. An error ε₄=N*-N is obtained by comparing the speed command value N* with a speed value detected N in the comparator C₄. The error ε₄ is input to the speed control compensating circuit G N (S) to carry out proportional amplification or integral amplification. Thus, the amplitude command value I m of the load current I L to be supplied to the motor winding is determined. The multiplier ML is operative to multiply the amplitude command value I m by a unit sinewave (one phase) sinωt synchronous with a rotational position of the motor. The output I L *=I m ·sinωt serves as a current command value which will be supplied to the armature winding (one phase) of the motor. Where a three-phase motor is employed as a load, current command values for respective phases are expressed as follows: I* LU = I m ·sinωt I* LV = I m ·sin(ωt-π/3) I* LW = I m ·sin(ωt-2π/3), where ω represents a rotational angle frequency of the motor. When N* > N, the error ε₄ becomes a positive value, the amplitude command value I m is increased to increase a load current (armature current), thus allowing a torque produced to be increased. Accordingly, the motor is accelerated, with the result that the relationship N*=N is eventually established, thus placing the system in a balanced condition. In contrast, when N* < N, the error ε₄ becomes a negative value to allow the amplitude command value I m to be decreased or to be negative, thus decreasing a torque produced or by applying a regenerative braking to the motor. As a result, the motor is decelerated with the result the relationship N*=N is also eventually established, thus placing the system in a balanced condition. Where the cycloconverter according to the present invention operates under condition of a load current below the rated value, it effects a control such that the reactive power at the receiving end becomes zero. However, where the cycloconverter operates under an overload condition, it effects a control such that the command value Q₃* for the reactive power control at the receiving-end is changed depending upon the magnitude (amplitude I m ) of the load current. Namely, in Fig. 5, the current amplitude command value I m is input to the reactive power command value circuit FQ through the absolute value circuit ABS, thus obtaining the reactive power command value Q S * at the receiving-end. Fig. 6 is circuit diagram illustrating an example of the reactive power command value circuit FQ wherein OA₁ and OA₂ represent inverting operational amplifiers, R₁ to R₅ resistors, D a diode, and VR a level setter. By using the level setter VR, an rated value I mo of the load current is set. Assuming that the resistance values of the resistors R₁ to R₃ provided in the inverting operational amplifier OA₁ are equal to each other, (I mo -|I m |) can be determined as its output. When I mo -I m , i.e., a load current is less than the rated value, the value of (I mo -|I m |) becomes positive. However, the output of the inverting operational amplifier OA₁ is forcedly suppressed to be zero by the action of the diode. In contrast, the value of (I mo -|I m |) is output from the inverting operational amplifier OA₁. On the other hand, the inverting operational amplifier OA₂ has an amplification factor K Q =(R₅/R₄) . Accordingly, when |I m [≦I mo , the reactive power command value Q S * becomes equal to zero. In contrast, when |I m |>I mo , the reactive power command value Q S * is expressed as Q S *=K O (|I m |-I mo ) . Fig. 7 shows the above-mentioned relationship between I O * and Q S *; wherein the load current amplitude |I m | of the reactive power command value Q S * is always set at zero until reaching the rated value I mo , and increases in proportion to the amplitude |I m | under an overload operation where a load current is above the rated value I mo . In this instance, a proportional constant K Q is determined as a substantially constant such that the circulating current I O of the cycloconverter is not interrupted. Accordingly, the circulating current I O of the cycloconverter decreases in inverse proportion to the absolute value I m of the amplitude of the load current I L so that the reactive power Q S becomes zero until the absolute value I m becomes equal to the rated value. In contrast, when the cycloconverter is operated under condition the absolute value I m exceeds the rated value, the reactive power Q S at the receiving-end does not become zero, thus shifting to a control such that the circulating current I O is not interrupted in accordance with the lagging reactive power. It is to be noted that a control is not carried out such that the circulating current I O under the overload operation is completely kept constant, but a control is effected so that the reactive power at the receiving-end becomes equal to its command value Q S *. Accordingly, from a microscopic point of view, the circulating current chances every moment, thus maintaining the reactive power Q S at a constant. Such a control provides an good effect with respect to harmonics of an input current of the cycloconverter. Namely, where an ac load is employed, there appear harmonics relevant to a frequency on the side of the load in the input current of the cycloconverter. It is considered that occurrence of such harmonics is due to changes in an effective power at the receiving-end of the cycloconverter and changes a reactive power at the receiving-end of the cycloconverter. In the case of a three-phase balanced load the effective power becomes constant, with the result that harmonics of an input current caused thereby do not occur. However, the reactive power at the receiving-end of the cycloconverter changes every moment according to a phase control. Accordingly, even in the three-phase balanced load, there is a possibility that harmonics occur in a current on the input side. By maintaining the reactive power at the receiving-end of the cycloconverter at a constant, it is possible to decrease harmonics (particularly, sideband waves coupled to a fundamental wave) on the input side. The cycloconverter system according to the present invention effects a control such that the reactive power is kept constant, i.e., Q S =0 until the output reaches the rated load, thus making it possible to reduce harmonics on the input side for the reason stated above. Further, the cycloconverter according to the present invention effects a control such that the reactive power is kept constant, i.e., Q S =Q S * from a microscopic point of view under an operation above the rated load, thus making it possible to reduce harmonics in a current on the input side. In the above-mentioned embodiments, it has been described that the single-phase load is employed. However, the present invention can be practiced in connection with a two-phase load or a polyphase load in the same manner as stated above. Further, the present invention is also applicable to a cycloconverter delta-connected. Furthermore, when a polyphase output cycloconverter is employed, distribution of circulating currents in respective phases is possible. In addition, it is needless to say that various applications can be possible within a range where the gist of the present invention is not changed. As stated above, the present invention can maintain the advantageous features of a cycloconverter of the circulating current type (small distortion in an output current, and high upper limit of an output frequency) even in an overload operation, thus ensuring a stabilized operation. Further, the present invention makes it possible to always maintain a power factor at the receiving-end at 1 until the output reaches the rated load operation and to have no necessity of remarkably increasing the capacitance of a power factor compensating capacitor connected at the receiving-end of the cycloconverter. Accordingly, an increase in the capacity of a power transformer or a converter required for an overload operation is relatively small, thus an overload operation is relatively small, thus enabling an operation with a high efficiency. In addition, even in an overload operation, only the reactive power command value Q S * at the receiving-end changes and the reactive power control is still continued. Namely, from a microscopic point of view, a control is maintained such that the reactive power is constant. Accordingly, this makes it possible to reduce harmonics (particularly, sideband waves coupled to a fundamental wave) on the input side which would occur due to changes in the reactive power.

Claims: Cycloconverter apparatus comprising: a cycloconverter ( CC ) of the circulating current type for connection to a single or poly-phase load, said cycloconverter ( CC ) being adapted to receive power from an AC power supply; a power factor compensating capacitor (CAP) connected to a receiving end of said cycloconverter ( CC ); means (CT S ,PT S ,VAR) for determining a reactive power (Q,Q S ) of the cycloconverter apparatus; a command value generator (C₁,H Q (S);C₄,G N (S),ABS) connected to the means for determining the reactive power and responsive to the reactive power (Q,Q S ) determined for generating a command value (I o *); comparator means (C₂) connected to the command value generator for comparing the circulating current (I o ) flowing in the cycloconverter ( CC ) with the command value (I o *) generated; and controlling means (G o (S),A₂,A₃,PHN,PHP,A₁) connected to the comparator means (C₂) for controlling the circulating current (I o ) flowing in said cycloconverter ( CC ) in response to the result of the comparison by the comparator means (C₂) characterised by further comprising: command value adjustment means (LIM,A₁,VR o ;FQ) connected to the command value generator for changing the command value (I o *) generated by said command value generator when an output current (I L ) of said cycloconverter ( CC ) is greater than a rated value such that the command value (I o *) is set and remains substantially constant regardless of variations in output current (I L ) greater than the rated value whereby the circulating current (I o ) flowing in said cycloconverter ( CC ) remains substantially constant. Cycloconverter apparatus according to claim 1, characterised in that said command value generator (C₁,H Q (S)) is also responsive to a reactive command value (Q S *) and said command value adjustment means (FQ) outputs a reactive command value (Q S *) of zero when said output current (I L ) is less than said rated value and outputs a reactive command value (Q S *) which increases in proportion to the magnitude of the output current (I L ) when the output current (I L ) is above said rated value. A control method for a cycloconverter of the circulating current type according to claim 1 comprising the steps of: generating a circulating current command value (I o *) in response to a reactive power (Q,Q S ) at a receiving end of the cycloconverter ( CC ); comparing a circulating current (I o ) with a circulating current command value (I o *); and controlling the circulating current (I o ) in response to the result of the comparison whereby a lagging reactive power of the cycloconverter ( CC ) and a leading reactive power of the power factor compensating capacitor (CAP) connected to a receiving end of the cycloconverter ( CC ) are cancelled with each other when an output current (I L ) of the cycloconverter ( CC ) is less than a rated value and characterised in that, when the output current (I L ) is greater than the rated value, the circulating current command value (I o *) is set and remains substantially constant by operation of command value adjustment means (LIM,A₁,VR o ;FQ) regardless of variation in the output current (I L ) greater than the rated value whereby the circulating current (I o ) remains substantially constant.

Inventors: TANAKA, SHIGERU
Applicants: KABUSHIKI KAISHA TOSHIBA
Application year: 1985
Main classification: H02M5
IPC codes: H02M7, H02M5