A COMPOUND DIELECTRIC MULTI-CONDUCTOR TRANSMISSION LINE AND DEVICES CONSTRUCTED THEREFROM

Abstract: A transmission line comprising a multi-layer dielectric slab structure including: a dielectric substrate layer (30) having a thickness ds and permittivity epsilon s; a conductive ground plane (31) on the bottom surface of the dielectric substrate layer (30); a dielectric guiding layer (32) having a thickness h and permittivity epsilon g, where epsilon g > epsilon s, attached to the top surface of dielectric substrate layer (30); at least one elongated and relatively narrow dielectric loading strip layer (33) having a width W, thickness dl, and permittivity epsilon l, where epsilon g > epsilon l, attached to the top surface of the dielectric guiding layer (32); and a conductive coating (34) on the top surface of the dielectric loading strip layer (32). Such a structure permits single mode propagation over a relatively wide band. Radiation losses due to coupling of the desired mode to the substrate modes and the conductors are furthermore reduced and the polarization of the dominant mode is such as to render said structure relatively insensitive to small deviations from parallelism among the different interfaces.

Description: A COMPOUND DIELECTRIC MULTI-CONDUCTOR TRANSMISSION LINE AND DEVICES CONSTRUCTED THEREFROM Background of the Invention This invention relates generally to high frequency transmission lines and more particularly to planar type waveguide structures for millimeter wave •applications. Guided wave transmission lines are widely used to channel the flow of high-frequency electrical energy. Common examples of these transmission lines are the coaxial line, the hollow metallic waveguide and the optical fiber. All these waveguiding structures are useful in long link applications. However, in situations where the distance between the transmitting and receiving points is below a few inches, as in an integrated cir¬ cuit, planar transmission lines offer an attractive alternative to these types of transmission lines. A variety of planar transmission line configurations are possible. In one type of planar transmission line, metallic conductors play a primary role in the waveguid¬ ing process. This includes the well known microstrip transmission line, the slotline, the coplanar waveguide, and coplanar strip line. In another type of planar transmission line, a dielectric strip plays a primary role in the waveguiding process. This includes the well known dielectric strip guide and the inverted strip guide, typical examples of which are respectively disclosed in U.S. Pat. 4,028,643, entitled "Waveguide Having Strip Dielectric Structure", which issued to T. Itoh on June 7, 1977 and U.S. Pat. 4,463,330, entitled "Dielectric Waveguide" which issued to T. Yoneyama on July 31, 1984. In general, all of these offer significant savings in size and weight over the non-planar varieties. Further, monolithic and hybrid technologies are closely compatible with the planar configuration. Consequently, these technologies can be used to generate with higher producibility, systems which offer superior performance and enhanced reliability. When combined with high- volume batch fabrication, significant cost savings can result. Summary This invention is directed to a multi-layer dielectric slab structure including a substrate layer and a guiding layer, with the substrate layer being bounded on the bottom side by a metal ground plane. A dielectric loading strip metallized on the top face comprises the top layer and forms the remainder of the structure. The layers of dielectric are primarily chosen with respect to their permittivities to keep the propagating energy away from the conductor surfaces, and thereby reduce conductor losses. The conductive coating on the top surface of the dielectric loading strip permits single mode propa¬ gation over.a relatively wide band and radiation losses due to coupling of the desired mode to the substrate modes are reduced and the polarization of the dominant mode is such as to render the structure of the trans- mission line relatively insensitive to small deviations from parallelism among the different interfaces of the structure. A whole family of devices evolves from such a structure. These include antennas, power couplers, switches, isolators and circulators. Brief Description of the Drawings Figure 1A shows a perspective view of a known prior art microstrip line; Figure IB shows a perspective view of a known prior art slotline; Figure 1C shows a perspective view of a known prior art coplanar waveguide; Figure ID shows a perspective view of a known prior art coplanar strip; Figure 2A shows a perspective view of a known prior art dielectric strip guide; Figure 2B shows a perspective view of a known prior art inverted dielectric strip guide; Figure 3 is schematically illustrative of a guided wave mode in a slab type waveguide in accordance with the subject invention; Figure 4 shows a perspective view of one embodiment of a compound dielectric multi-conductor transmission line in accordance with this invention; Figure 5 shows a perspective view of an alter¬ native embodiment of the transmission line shown in Figure 4; Figure 6 is a perspective view illustrative of a first embodiment of an antenna constructed in accordance with the waveguide structure shown in Figure 4; Figure 7 is a cross-sectional view of the antenna shown in Figure 6 taken along the lines 7-7 thereof and further illustrates the energy distribution across the interface of the dielectric layers to cause radiation; Figure 8 is a perspective view illustrative of a second embodiment of the antenna configured from the transmission line shown in Figure 5; Figure 9 is a perspective view illustrative of a first embodiment of an isolator device configured from the transmission line shown in Figure 4; Figure 10 is a perspective view illustrative of a second embodiment of an isolator device constructed from the embodiment of the transmission line shown in Figure 4; Figures 11A and 11B are schematic diagrams helpful in understanding the operation of the isolator device as shown in Figures 9 and 10; Figure 12 is a perspective view illustrative of a circulator type of coupling device configured from the transmission line shown in Figure 4; Figure 13 is a schematic diagram illustrative of the pattern of the top loading strips formed on the device shown in Figure 12; Figure 14 is a partial cross sectional view illustrative of a modification of the embodiment shown in Figure 12; Figure ^ .15 is a perspective view illustrative of an optically controlled switch device formed in accordance with the transmission line shown in Figure 4; Figure 16 is a perspective view illustrative of a device for providing a transition from rectangular waveguide to the subject transmission line as shown in Figure 4; Figures 17 and 18 are schematic diagrams helpful in understanding the operation of the transi¬ tion device shown in Figure 16; Figure 19 is a perspective view illustrative of an embodiment for providing efficient coupling of power generated in an external device into the waveguide structure as shown in Figure 4; and Figure 20 is a perspective view illustrative of another embodiment of a configuration of coupling power generated in an external device into the trans- mission line structure shown in Figure 4. Detailed Description of the Invention Before considering the preferred embodiments of the invention, the details and limitations of known prior art structures shown in Figures 1A through ID and 2A and 2B will first be discussed. The development of planar waveguiding struc¬ tures for millimeter (mm)-wave applications has been proceeding for about two decades. An important concept in propagating energy in a waveguide is the notion of a mode. A mode is a spatial distribution of energy across the cross-section of the guiding structure. In general, a waveguiding structure can propagate several modes. Each of the modes has a characteristic cut-off fre¬ quency below which the waveguiding structure will not support it. It is customary to choose the cross sec¬ tional dimensions of the waveguiding structure such that, over the frequency range of interest, only one mode will be * 'supported. This mode is often the one with the lowest cut-off frequency and is called the dominant mode and the cut-off frequency for the next higher order mode represents the useful bandwidth for the wave¬ guiding structure. Accordingly, it is customary to design structures for the widest possible bandwidth consistent with single-mode operation as described above. An important consideration in the design of a planar structure is the nature and behavior of the so- called substrate modes. These are undesirable parasitic modes which, if allowed to propagate, can cause severe transmission losses, especially at waveguide bends and discontinuities. Such consideration must be given attention over a wide range of geometric parameters, as well as over the entire operating frequency range. For example, in the non-radiative waveguide disclosed in the above referenced U.S. Pat. 4,463,330, Yoneyama, it is now established that small gaps, due to fabrication imperfections, between the metal planes and the dielectric materials leads to radiation loss. This occurs primarily because the electric field for the dominant mode is sub¬ stantially parallel to the metal planes. It is desir¬ able to establish the polarization of the dominant mode such that the electric field is largely normal to the metal planes. This is not possible with the non- radiative guide. Similarly, the strip dielectric waveguide structure of Itoh, disclosed in the above referenced U.S. Pat. 4,028,643, also has limitations. For one, its dominant mode is actually an admixture of two orthogonal polarization states. Consequently, some leakage via coupling to the parasitic substrate modes is unavoidable. Further, this coupling grows stronger as one progresses to lower frequencies since then, the difference between the guiding structure and the region external to it cannot be distinguished by the propagating energy. Additionally, the struc-ure has no inherent mechanism for propagating direct current energy. Considering now the drawings, in the structures shown in Figure 1A through ID, which are illustrative of known prior art, metallic strips are of primary importance in the waveguiding process. In the microstrip line of Figure 1A, a conducting strip 10 is mounted on a dielectric 11 which is coated on its bottom surface with a metallic ground plane 12. In the slotline of Figure IB, two parallel conductors 13 are placed upon dielectric 14. In Figure 1C, a coplanar waveguide configuration is shown in which three parallel con- ductors 15 are placed on dielectric 16, the two outer conductive strips acting as a ground plane. In Figure ID, coplanar conductive strips 17 are mounted on dielectric 18, but the edges of the strips are not coextensive with the edges of the dielectric slab, as they were in the slotline of Figure IB. The microstrip structure shown in Figure 1A has proven the most versatile and successful among the prior art configurations using metallic strips. The microstrip type of transmission line has been success¬ fully used in applications up to 60GHz, but even at those frequencies, some of the problems associated with its use are evident. In microstrip, the substrate modes are suppressed by choosing a dielectric substrate that is thin enough. At 60 GHz, a typical substrate thickness must not exceed 8 mils. At higher frequencies, even thinner substrates must be used. The requirement of thin substrates bears important consequences for the electrical and mechanical properties of the microstrip structure. The impedance of a transmission line in microstrip is primarily determined by the ratio of the conductor strip width W, to the dielectric thickness h, i.e. w/h. The value of W is bounded at the upper end by the requirement that it be small compared to the wavelength of the propagat¬ ing energy at the frequency in question. The lower bound on W is determined by the accuracy and reproduc- ibility with which a narrow line can be fabricated. These bounds in turn limit the range of line impedances available to the circuit designer. Consequently, the versatility of the structure is limited when very thin dielectric substrates are used. Another serious problem is transmission line loss. In microstrip, this loss is dominated by the ohmic losses in the metal conductors. These losses inherently increase with frequency, and in microstrip, are made to increase even more rapidly as thinner di¬ electric substrates are used. A third problem is related to fabrication. The thin substrates required make for very delicate in-process handling. Such processing conditions can result in poor fabrication yields. A fourth problem concerns the thermal pro¬ perties of the structure. Ironically, this consideration leads to the conclusion that the substrate is not thin enough. If the dielectric substrate is a semiconductor on which truly planar transmitting sources are inte¬ grated, then the heat generated within these sources would have to be removed if the device is to survive operation. Unfortunately, most electrical insulators are also thermal insulators, diamond and beryllium oxide being exceptions, and consequently, the heat generating device would be thermally isolated from a heat sink, unless the substrate was made very thin. Several attempts have been made to reduce the impact of the above drawbacks, by inserting these sub¬ strates into waveguide enclosures to form fin-line and suspended-substrate-stripline configurations. However, these structures are limited by the dimensions of the • waveguide enclosures in which they are housed. Besides, the advantages of size and weight are compromised somewhat, and the thermal problem is left unaddressed. The shortcomings of microstrip are described in Pucel, R.A., "Design Considerations for Monolithic Microwave Circuits", IEEE Trans., Vol. MTT-29, no. 6, pp. 513- 534, June, 1981. Planar dielectric waveguides, on the other hand, offer more convenient substrate and guide dimen- sions, and also have low loss. Considering now Figures 2A and 2B, in Figure 2A, a dielectric strip guide is shown in which a dielectric strip 19 is mounted on a dielectric slab 20 which is coated on its bottom surface with metallic ground plane 21. An inverted strip guide is shown in Figure 2B in which a dielectric strip 22 is sandwiched between dielectric slab 23 and metallic ground plane 24. A key feature " of planar dielectric structures is that they have very low loss at frequencies where the structures of Figure lA-Figure ID cannot be used at all. Thus, planar dielectric waveguides have been used at optical frequencies spanning the infrared to visible range. This is in part due to the outright absence of conductors or the relative remoteness of the conductor surfaces from the propagating energy. An elementary way of perceiving the guiding process is illustrated in Figure 3. If light in an optically denser medium is incident on an interface with a relatively rarer medium, then total internal reflection off the interface occurs whenever the angle of incidence Q in the denser medium exceeds a certain critical angle. This critical angle is characteristic of the pair of materials forming the interface. If a slab of optically dense material, e.g. glass is sand¬ wiched by optically rarer medium e.g. air, then wave¬ guiding is possible by total internal reflection off both interfaces. The optically dense medium is called the guiding layer; the bounding rarer medium is called the cladding layer. The structure is appropriately called a slab waveguide. The planar dielectric structures of Figures 2A and 2B, however, suffer from the inherent limitation of being multi-modal in that they all support at least two modes. Any attempt to realize single mode operation usually results in a mode that is too weakly bound to the structure to be of any practical use. The very close separation in the cut-off frequencies between the dominant T Q mode and the next higher TE mode forces either an acceptance of a very narrow band waveguiding structure, or a dual-mode guide. The coupling between these two modes can result in high radiation loss at discontinuities and bends, as well as increased coupling to the spurious substrate modes mentioned in connection with microstrip and dielectric strip waveguide. A second disadvantage of planar dielectric structures is their extreme sensitivity to the condition of the interface. Thus, any roughness in the surfaces of the guiding or cladding media or any bubbles trapped between them during the bonding process can have a profound influence on the losses due to random scatter¬ ing from these centers at the boundaries. In view of the foregoing, the details of the various embodiments of this invention will now be considered. Referring now to Figure 4, a substrate 30 whose permittivity is €s and whose thickness is ds is clad on one side, i.e. the bottom face, by a metal ground layer 31. The other side or top face of the substrate 30 is bonded to a dielectric guiding slab layer 32 whose permittivity is <= and whose thickness y is h. A relatively narrower dielectric loading strip 33 of width W, thickness d, and permittivity 6, is bonded to the other side or top face of guiding slab 32. The propagating direction for the electrical energy is along its longitudinal axis. The upper face of the loading strip 33 is clad with a relatively thin metal layer or coating 34, which covers at least one third of the width W of the loading strip and extends uniformly, periodically or aperiodically along its length depending on the needs of the user. The permittivity £ of the guiding layer 32 is made to be greater than both the permittivity S of the substrate 30 and the permittivity £ . of strip 33 due to the fact that energy being propagated seeks the medium with the highest permittivity. The nature and thickness of metal ground layer 31 and metal cladding layer 34 are not critical. A waveguide structure in accordance with the configuration shown in Figure 4 was constructed with a guiding slab layer 32 of RT "Duroid" 6010 and both the substrate dielectric 30 and strip 33 of alumina. "Duroid" is a trademark of Rogers Corp. for filled tetrafluoroe thylene material. The permittivities and dimensions were as follows: € g = 10.6 € Q € 1 - " 8 = 9 ' 7 * 0 where 0 n is the permittivity of free space, and h = 0.025" d,^ = 0.020" 1 s When the permittivities of the substrate 30 and strip 33 are equal, €, = 6 the preferred ratio between each of the thicknesses ds and di, of the substrate and strip and that of the guiding slab layer h is approxi¬ mately 0.75. In the example described above, for practical reasons the thickness ratio was altered to 0.8. The line loss in this waveguide was measured at 94GHz and was found to be only 0.4db/inch, compared with a loss of 2.5db/inch at that frequency for micro- strip, an improvement of almost six to one. The transmission line of Figure 4 combines the wideband feature of microstrip and the low loss characteristic of planar dielectric waveguides. Like the planar dielectric waveguide, it has no "sidewalls" so that scattering losses are reduced. Ohmic conductor losses are reduced substantially below those in an equivalent microstrip structure, permitting operation at higher frequencies. Further, the amount by which they are reduced increases as the frequency is increased, In the example presented, the conductor losses are 56% of their microstrip contributions at 75GHz. At 100GHz, they are 33% of their microstrip contributions. This is a significant result since conductor losses are known to increase with increasing frequency. It was earlier remarked that in the planar dielectric waveguide, including the strip-loaded guide, the dominant TM Q mode is not widely separated from the TE Q mode. The addition of metal conductors 31 and 34 in the manner shown is particularly significant in the instant invention in that the separation between these modes is widened, thus permitting single-mode operation over a wider band. The actual separation is determined by the ratio h/(d +d,) i.e., a larger ratio results in a wider separation between the modes and hence, a wider operating bandwidth independent of the dielectric materials chosen so long as = G . The condition for mode separation is somewhat more complicated but nevertheless calculable when € ≠ <=, . In either case the dominant mode is the TM Q mode whose polarization is such that, the electric field is largely at right angles to the metal conductors as required. This entire phenomenon was not previously anticipated. Also, the top conducting strip 34 serves as a reflector to confine the propagating energy to region 1 of the dielectric layers even at low frequencies. The substrate modes in the "wings" of the structure (region 2) have no such confinement means. Consequently, the effective dielectric constant of the wings will decrease with frequency whereas it will remain substantially constant in the region under the strip. This difference in the effective dielectric constants is what produces the confinement of the propagating energy which will there¬ fore be well guided even at low frequencies and dc. The thickness of the microstrip substrate 11 of the known prior structure shown in Figure 1 is limited because of the need to suppress the spurious substrate modes, as indicated previously. The most troublesome among these modes is the TE n mode for a grounded dielectric slab. The structure shown in Figure 4, on the other hand, suppresses the propagation of this TE Q mode, thus permitting the use of thicker substrates 30 at a given operating frequency than would be possible with a comparable microstrip guide. As a result, losses at waveguide bends and waveguide dis¬ continuities will be greatly reduced. As illustrated by the foregoing example, the dimensions of the waveguide (Figure 4) at the frequency range of 75-100GHz will be larger than those of a micro¬ strip structure (Figure 1) designed for that range. The choices of the guiding layer 32 thickness h, the substrate dielectric 30 thickness ds, and the loading strip 33 thickness d,, are determined by the desired frequency of operation and by the dielectric permittiv- ities € , €_ and €, . At a given frequency, a larger difference £g - £s or £g - £,l will lead to smaller values of h, d S and d,JL, respectively. For example, if 6S - €. = 6.6 £ 0 which corresponds to BeO, and £ = 12.9 € Q which corresponds to GaAs.then h = 0.012" and d = d. = 0.010" in order to yield a waveguiding structure of identical performance to the one quoted in the foregoing example. Thus, the advantage of thicker substrate material is surrendered somewhat if a large dielectric discontinuity exists at the relevant interfaces. Any or all of the dielectric elements 30, 32 and 33 of the structure shown in Figure 4 may be semi¬ conductors. Another unexpected result that emerges from a consideration of the new structure, particularly when, semiconductors are used, is a method for exciting the dominant mode. The conventional shunt and series excitation in microstrip line are well known. In the present guide of the present invention, however, another excitation method exists. The excitation source may be located at the interface 29 between guiding layer 32 and substrate dielectric 30 or at the interface 31 between guiding layer 32 and strip 33. The excitation source would be oriented with its current transport direction parallel to the desired direction of propa¬ gation of the energy i.e. parallel to the longitudinal axis of the strip. This method of excitation is useful because: (i) the interface 29 or 31 is the natural location of a device integrated on a semiconductor guiding layer 32; and (ii) substrate 30 and strip 33 would provide dc isolation of transmission line conductors 31 and 34 from such a device. This is a convenient feature which adds to the design flexibility of the structure. In many respects, the transmission line of Figure 4 resembles microstrip with the closeness of the resemblance under the designer's control. At low frequencies, its behavior is identical to microstrip. In effect, the structure of the present invention may be viewed as a means to extend the frequency of opera¬ tion of microstrip circuits without having to change the substrate thickness. Thus, a 70 mil thick conven¬ tional microstrip configuration is only usable from dc to 14GHz while the 70 mil compound dielectric slab in accordance with the invention and as presented in the design example above is usable from dc to 100GHz. The .characteristic impedance of the trans¬ mission line shown in Figure 4 is determined primarily by the ratio of the width W of the loading strip 33 to the effective guiding layer thickness, (which is always somewhat larger than the actual thickness h) , when the width is small compared to the wavelength. Changes in width W can also be used to provide impedance matching and frequency filtering. For widths comparable to or larger than the wavelength, no complicated field analysis is required to define the impedance level, which is dependent on the operating frequency. However, this change need not be very large. In the design example presented, a 50Λ. line at 75GHz becomes a 64-Λ. line at 100GHz. Smaller variations in impedance are possible with alternative designs at the cost of higher line loss. In the structure of Figure , an appreciable amount of energy is propagated in the guiding and the substrate layers 32 and 30 where no "sidewall" losses are manifested. A smaller proportion of energy is present in the loading strip 33. This energy, however, is subjected to sidewall scattering loss. Besides, the fields at the edge of the conducting strip 34 are relatively higher, so that the scattering losses could be greater. A solution to this problem is to taper the sides of the dielectric loading strip 33 so that its cross section is no longer rectangular. Figure 5 shows the special case of a symmetric linear taper so that loading strip 36 has the cross section of an isoceles trapezoid. The rest of the configuration is similar to that previously described with respect to Figure 4. Strip 36 rests on guiding slab layer 37 which is in turn mounted on substrate dielectric layer 38. Ground plane 39 is coated on the bottom of layer 38 while a metal cladding layer 40 is formed on the top surface of the taper.ed loading strip 36. A variety of other tapers may also be used for strip 36 such as concave and convex circular, concave and convex hyperbolic, exponential, etc. The technique of tapering the strip 36 has some additional latent advantages: (i) It permits a wider range of conductor linewidths and can be realized without running into the mechanical difficulty of having to mount very thin strips edge-on on the guiding layer; (ii) The tapered sides "soften" the discontinuity at the strip's edge. This has the effect of focusing the energy towards the center of the strip. This focusing effect is increased if the taper is such that the slope at any point on it relative to the vertical is greater than the critical angle for that interface. The tapered sides also increase the separation between the TE Q and TM Q modes beyond the effect previously described. Thus, a wider operating bandwidth is permitted. Alternatively, for a given operating bandwidth, the conductor losses may be reduced even further. One of the anticipated disadvantages is the sensitivity of the propagating energy to imperfections of the tapered sides. This is expected to be more critical for high impedance (narrow conductor width) lines. However, such sensitivity will have a smaller impact than any corresponding effect in a competitive planar dielectric waveguide. The wide bandwidth afforded by the waveguide of this invention makes the medium ideally suited for digital transmission. Additionally, one or more of the dielectric layers 30, 32, 33 of Figure 4 may be replaced by a non-reciprocal medium, including, for example, a ferroelectric or ferrimagnetic material such as Barium titanate or a ferrite. The relatively small volumes in which the propagating waves are confined would enable one to use smaller amounts of control energy, and yet maintain the control energy density (energy/volume) at high enough levels to manipulate the guided energy. In practice, this means that one may use smaller mag¬ netic field strengths to manipulate the high-frequency energy in devices such as ferrite phase shifters and modulators as well as in circulators and isolators. The heat dissipation problem outlined earlier can be more effectively overcome by using materials, such as BeO, that are electrical insulators but thermal conductors for the substrate dielectric and/or dielec- trie strip. Since these materials can be brought in direct contact with the power-generating device, they can serve as a low thermal resistance path between the device and a heat sink. This now leads to a consideration of the family of devices shown in Figure 6 through 20. Antenna arrays are widely used in microwave and millimeter wave ground based and airborne systems. Planar arrays are attractive for airborne applications because of their lightweight structural compatability and low cost. Accordingly, Figures 6 and 8 are illus¬ trative of two embodiments of antennas which can be configured from the basic transmission line structure shown in Figure 4. As shown in Figure 6, the dielectric loading strip 33 on top of the guiding layer 32 includes an abrupt transition to a region 42 of reduced thickness a which includes a plurality of transverse metal lines or strips 44 formed on the top surface thereof and which operate as a diffraction grating. Due to the fact that energy propagated in the guiding layer 32 has a dis¬ tribution pattern across its thickness as shown by reference numeral 46 of Figure 7, part of the energy couples to the adjoining regions of the loading strip 33 and the substrate 30, which if the thickness dimension is properly chosen, then sufficient energy exists in the region 42 to be radiated from the diffraction grating in a well known manner. While the transition in the loading strip 33 of Figure 6 is shown as a step, the transition, when desirable, can be sloped and/or the grating element 44 can be formed in different shapes with the width and spacing of these elements being either periodic or aperiodic depending upon the desired radiation pattern. Also, the reduced region 42 can be made of a different dielectric which leads to consideration of the embodiment shown in Figure 8. There a linear tapered transition of both the dielectric loading strip 33 as well as the top layer of metallization 34 is provided as shown by reference numerals 48 and 50, respectively, with a pointed radiator element comprised of dielectric having a permittivity € , being provided as an exten- sion of the layer 33. The permittivity <= ■ , of the radiating element 52 is made to be greater than the oermittivities of both the guiding layer 32 as well as the permittivities £ and C. of the layers 30 and 33. Since energy propagating in the transmission line seeks the highest dielectric medium, the radiating element 52 in effect lifts the guided mode off of the guiding layer 32. The radiating element 52 as shown also includes a tapered nose section 54 so as to efficiently radiate energy to free space or alternatively, to couple energy efficiently from free space into the transmission line. When desirable, the element 52 can be configured to include a suitable corrugation grid pattern or a combination thereof in a manner shown in Figure 6. It may also be possible to fabricate the radiating element 52 from materials which are alterable electronically so as to alter the direction of radiation and thereby provide a scanning mode. Turning attention now to the embodiments shown in Figures 9 and 10, shown thereat are two configura¬ tions of a device known as a signal isolator which permits the flow of a signal on one direction but not in the other direction. This is provided by the inclusion of a section of an anisotropic material such as a ferrite or electro-optic medium into the transmission line structure of Figure 4. As shown in Figure 9, a section 56 of magnetically biased ferrite which is tapered at both ends is inserted in the length of dielectric loading strip 33 beneath the outer layer of metallization 34. The taper is for providing an im¬ pedance match between ferrite material of the section 56 and the dielectric material of the loading strip 33. A generally rectangular slab 58 of lossy absorber material is affixed to one side 60 of the ferrite section 56, the purpose of which will be considered subsequently. A second embodiment of an isolator device is shown in Figure 10 and differs from the embodiment shown in Figure 9 in the shape of the ferrite section included along the length of the dielectric strip 33. As shown, a section 56' of ferrite material having a sloped type of transition is included. These devices operate in the following manner. For isotropic material, under normal operation, the electromagnetic field is distributed symmetrically with respect to the cross section of any transmission line. However, when an anisotropic material is inserted into the structure and is suitably biased by a magnetic field Hα,c, the field crowds towards one edge of the line for a wave propagating in the guiding layer 32 in one direction, but crowds towards the other edge when the direction of propagation is reversed, as shown in Figures 11A and 11B. Accordingly, means for generating a magnetic biasing dc field H, and which may be com¬ prised of electromagnets, not shown, applies the magnetic field upwardly through the dielectric layers 30, 32 and 33 as well as the ferrite section 56 and 56' and the energy propagating along the line will be crowded towards the slab 58 of absorber material in one direction as shown in Figure 11 while being crowded toward the opposite edge for energy propagation in the opposite direction. Thus energy will be absorbed strongly in one direction of propagation and not so for the reverse direction, thus providing a device which is commonly known as an isolator. Additional impedance matching and bandwidth requirements may be met by including a stepped or gradual discontinuity in the height of the ferrite sections 56 and 56' relative to that of the dielectric layer 33. In view of the characteristic of anisotropic material, such as ferrite which when suitably biased by a magnetic field causes the propagating field to crowd at one edge of the line for a wave propagating in one direc¬ tion and crowds at the other edge when the direction of propagation is reversed, this asymmetric field crowding can also be used to realize a device which is known as a circulator. A four port circulator configured in accordance with the subject invention is shown in Figure 12. There a pair of ferrite strips 60 and 62 shaped in the form of back to back U-shaped elements and having outer layers of metallization 42 formed thereon are bonded to the upper surface of the dielectric layer 32 in relatively close proximity to each other. The ferrite strips 60 and 62 have a spacing P over a length L so that energy being guided beneath the strips 60 and 62 in the guiding layer 32 can be cross coupled. As schematically shown in Figure 13, the ferrite strip 60 as well as the underlying guiding layer 32 and substrate layer 30, not shown, terminate in upper and lower port signals 2 and 3. It is a well known property that waves traveling in one direction along a transmission line excite waves which then travel in the opposite direction in another line which is electromagnetically coupled thereto. If a biased aniso- tropic material is used, e.g. ferrite, the asymmetric field distribution produces strong coupling between the lines for signal waves traveling in only one direction. Thus in the configuration shown in Figure 12, for a magnetic biasing field H, in a first predetermined direction, it will cause a wave entering port 1 to travel along the guiding layer 32 under the ferrite strip 60 to be inwardly crowded adjacent the portion of the guiding layer underlying the strip 62 and will be coupled to port 2 if the dimensions P and L are adjusted properly. However, a wave entering port 2 and propa¬ gating along the guiding layer 32 will have its field crowded at the edge remote from strip 60 and energy will emerge at port 3. Similarly a wave entering port 3 will emerge at port 4 and that entering port 4 will emerge at port 1. Therefore, what is provided is a circulator which will act as a clockwise "turnstile" for energy. By reversing the polarity of H> , the direction of circulation is reversed. By including means, not shown, providing a gradual reversal of H, in response to a control signal a modulator results while an abrupt reversal, on the other hand, results in a switch. Thus the isolator embodiment (Figures 9 and 10) can with reversal of the biasing magnetic field become a single-pole, single-throw switch while the circula- tor (Figure 12) becomes a double-pole, double-throw . switch. In order to adjust the degree of coupling of energy propagating along the guiding layer 32 under¬ lying the parallel ferrite strips 60 and 62, one may add a contiguous section 64 of ferrite or a dielectric ' between the parallel segments 60 and 62 as shown in Figure 14. It should also be noted that semiconductor materials can be used as one of the layers such as the loading layer 33, in the transmission line structure shown in Figure 4. This now leads to a consideration of Figure 15 where there is disclosed an embodiment of an optically triggered switch device including a semi¬ conductor member as one of the elements between the outer metallized layers 31 and 42. As shown in Figure 15, a loading strip 66 of semiconductor material having dimensions similar to the dielectric loading strip 33 (Figure 4), is affixed to the top surface of the dielectric guiding layer 32; however, a gap 68 is formed in the top metallization 42 where light from a triggered light source, not shown, is directed to and impinges on the semiconductor strip at 69. Adjacent the semi¬ conductor loading strip 66 is a relatively short arm member 70 of semiconductor material also coated with a top layer of metallization 42 and which is terminated at one end in a lossy load member 72. The arm is placed adjacent the semiconductor member 66 on the input side of the gap 68 and the dielectric or the semiconductor strip 66 is selected such that without the metallization 42, the waveguide is effectively open, i.e.. cut-off to millimeter waves propagating in the guiding layer 32; however, when the gap 68 is illuminated with light of suitable wavelength, a hole-electron plasma is generated at the upper surface 69 which bridges the gap in the metallization 42 so that the waveguide is no longer cut off and transmits millimeter wave energy in the under¬ lying guiding layer 32 past the gap 68. In the absence of incident light at the gap 68, input millimeter wave energy is reflected back from the gap 68 where it is coupled to the side arm region including member 70 into the load 72 while the waveguide is in a cut-off state. Alternately, the semiconductor material of the strip 66 can be chosen so that there is no cut-off condition. In this instance, the intensity and/or position of the optical beam directed to the gap 68 is used to modulate the amplitude and/or the phase of the millimeter wave propagating beneath the strip 66. Thus semiconductor materials can be used as one or more of the layers. This even includes the guiding layer 32. Furthermore, active and passive sources can be integrated into the semiconductor. Active devices should be aligned so that their current path is colinear with the long axis of the transmission line in order that energy may be effectively coupled to and from the line. The waveguide structure of the subject invention thus has the potential for realizing complete circuit and system functions on a single semiconductor wafer; in other words, it is compatible with monolithic integration. Since it is necessary in some instances to couple energy to and from other types of transmission lines and/or external devices, transitions and/or coupl¬ ing devices are required for the type of transmission line structure shown in Figure 4. One such transition is shown in Figure 16 and comprises a transition for use in connection with rectangular waveguide. As shown in Figure 16, the flange 74 of a piece of rectangular waveguide 76 includes a rectangular aperture 78 which is adapted to receive an outwardly projecting tapered wedge section 80 of the guiding layer 32 which is designed to fit into the interior of the waveguide 76 via the aperture 78. The principle of energy coupling is furthermore illustrated in Figure 17. Referring now to Figure 17, if a ray designated by reference numeral 82 is incident at an interface 83 of two dielectric media 84 and 86 where the relative dielectric permittivity € • of one, for example, media 84 is greater than the other, shown as unity in Figure 17, then the ray 82, having an incident angle ψ. relative to the normal 88 of the interface will be refracted as the ray 85 at an angle and wherein ψ is less than ψ. according to the Snell formula: sin ψ i = sin ψ_ where the dielectric media 84 has a shape corresponding to the tapered wedde section 80 and consisting of two converging interface surfaces then the refracted ray 85 encounters a second interface 87 of the same two materials except that this interface is at an inclina¬ tion angle Q relative to the first interface 83. Con¬ sequently, the refracted ray 85 has an incident angle of Q + relative to the normal 89. Since permittivity of the medium 84 is known to be greater than 1 while the permittivity of the medium 86 is unity, total internal reflection as shown by the ray 91 will occur if: β + ψ r > θ c = Bi-r l ( i/~mf x ) where Qc is the critical ang 3 le. If ψTi. mm. denotes the smallest incident angle ψ. for which total internal reflection occurs, then for all angles greater than 9 , all incident energy will then be confined to the medium 84, which in the embodiment shown in Figure 16 comprises the guiding layer 32. Referring briefly to Figure 18, shown sche¬ matically is the propagation of energy in the rectangular waveguide 76 for the dominant mode. The positioning of the wedge 80 of the dielectric layer 32 is shown inserted therein such that its plane is parallel to the broadwalls of the rectangular waveguide 76. Consider- ing the discussion with respect to Figure 17, the energy propagating in the waveguide 17 will be launched onto the guiding layer 32 through the tapered section 80 since the dominant mode in rectangular waveguide may be considered as a superposition of two criss-crossing rays 82. The angles between these two rays change with the frequency and accordingly, the useful operating bandwidth defines a range of angles i . For a given dielectric, one can therefore define θ such that coupl¬ ing will occur for the smallest value of ψ.. Considering now the Figures 19 and 20, shown thereat are two embodiments for coupling power between external power source devices and the waveguide structure of the present invention. With reference to Figure 19, an active device 90 is shown mounted on one side surface of a heat sink 92. Coupling to the di¬ electric guiding layer 32 is provided by a right-angled strip of metallization 94 which is formed over the face 96 of the active device 90 and onto the top sur¬ face of the guiding layer 32 where it extends under the bottom face of the loading strip 33 and terminating in a taper 98 as shown to provide an impedance match. Addi¬ tionally, a bias filter comprised of a pattern 100 of metallization consisting of wide and narrow sections of metal are formed on the top surface of the guiding layer 32 which are coupled by conductor line segments 101 between the tip 98 of the metallization strip 94 and a bias voltage source, not shown, coupled to ter¬ minal 102. The tapered strip of metallization 94 in effect acts like a metallized antenna which is inserted into the interface 31 between the guiding layer 32 and the loading strip 33 to radiate energy into the guiding layer 32 of the transmission line in the desired mode. When desired, the metallization strip 94 can be con¬ figured into a grating to maximize coupling to the device. The embodiment shown in Figure 20, on the other hand, is illustrative of a configuration wherein the active device 90 is mounted on the top surface of the heat sink 92, in which case coupling between the active device 90 and the guiding dielectric layer 32 is made simply by a planar section of metallization 94' extending from the top surface 104 of the active device to the interface 31 between the dielectric layers 32 and 33. The metallization 94' additionally includes a tapered end section 106 over the active device 90. As in the embodiment shown in Figure 19, the bias filter 100 is again coupled between bias voltage terminal 102 and the tapered tip 98 of the metallisation strip 94' by segments of a line conductor 101. Thus what has been shown and described is a slab type of structure for a millimeter wave trans- mission line including a family of devices which is realizable therefrom. The examples described above are only illus¬ trative of the specific embodiments of the invention and accordingly various other modifications will suggest themselves to those skilled in the art. Therefore, all modifications, alterations and changes coming within the spirit and scope of the invention are herein meant to be included.

Claims: CLAIMS 1. A transmission line structure comprising: (a) a first dielectric layer of predetermined thickness d and permittivity € ,and having top and bottom parallel surfaces; (b) a conductive ground plane on the bottom surface of said first dielectric layer; (c) a second dielectric layer of predeter¬ mined thickness h and permittivity € , where > e , g g s and having top and bottom parallel surfaces and attached to said first dielectric layer with the bottom surface of said second dielectric layer attached to the top surface of said first dielectric layer; (d) at least one elongated and relatively narrow third dielectric layer of thickness d.. and permittivity £, , where ζ„ ~> (- -> , and having top and bottom parallel surfaces, said bottom surface being attached to the top surface of said second dielectric layer; and (e) a conductive coating on said top surface of said third dielectric layer, whereby single mode propagation is permitted over a relatively wide band and radiation losses due to coupling of the desired mode to the substrate modes and the conductors are reduced, and the polarization of the dominant mode is such as to render said structure relatively insensitive to small deviations from parallelism among the different inter¬ faces. 2. The transmission line structure of claim 1 wherein said first dielectric layer comprises a sub- strate layer, said second dielectric layer comprises a guiding slab layer coextensive with said substrate layer, and said third dielectric layer comprises a dielectric loading strip layer having a width substantially less than the width of said substrate and guiding layers. 3. The transmission line structure of claim 2 wherein said conductive coating at least partially covers said top surface of said dielectric loading strip layer. 4. The transmission line structure of claim 2 wherein said conductive coating covers at least one third of the width of said top surface of said loading strip layer. 5. The transmission line structure of claim 2 wherein said conductive coating extends uni¬ formly, periodically or aperiodically along its length. 6. The transmission line structure of claim 2 wherein at least one of said layers comprises a layer of semiconductor material. 7. The transmission line structure of claim 2 wherein at least one of said layers comprises a non-reciprocal medium. 8. The transmission line structure of claim 2 wherein at least one of said layers is addition- ally thermally conductive. 9. The transmission line structure of claim 2 wherein the width of said loading strip layer is selectively changed to provide impedance matching and frequency filtering. 10. The transmission line structure of claim 2 wherein said loading strip layer includes a pair of parallel side surfaces. 11. The transmission line structure of claim 2 wherein said loading strip layer includes a pair of non-parallel side surfaces. 12. The transmission line structure of claim 2 wherein said loading strip layer includes a pair of tapered side surfaces. 13. The transmission line structure of claim 2 wherein said loading strip layer includes a region at one terminal end of reduced thickness (a) whereby energy propagating along said guiding layer diffuses to said reduced thickness region and wherein said conductive coating at said region includes means for coupling to or from free space, thereby providing an antenna structure. 14. The transmission line structure of claim 13 wherein said loading strip layer includes an abrupt thickness transition to said region of reduced thickness. 15. The transmission line structure of claim 13 wherein said energy coupling means includes means for generating a radiation pattern. 16. The transmission line-structure of- claim 15 wherein said means for generating a radiation pattern comprises a diffraction grating formed of said conductive coating at said region of reduced thickness. 17. The transmission line structure of claim 16 wherein said diffraction grating comprises a plurality of conductive elements of predetermined shape, width and spacing. 18. The transmission line structure of claim 17 wherein said plurality of conductive elements comprises a plurality of conductive strips arranged in parallel transversely across the width of said loading strip layer. 19. The transmission line structure of claim 2 wherein said loading strip layer includes a region at one terminal end having a permittivity 6 rad' where ζ ~-aά > ~q ' € s and ~l' whereb Y energy propagating along said guiding layer is substantially coupled to said end region, said end region including means for coupling energy to or from free space, thereby providing an antenna structure. 20. The transmission line structure of claim 19 wherein said energy coupling means comprises a tapered nose section. 21. The transmission line structure of claim 20 wherein said tapered nose section extends beyond the ends of the underlying guiding and substrate layers. 22. The transmission line structure of claim 19 wherein said end region is devoid of said con¬ ductive coating. 23. The transmission line structure of claim 19 wherein said end region includes means for generating a radiation pattern. 24. The transmission line structure of claim 19 wherein said end region is comprised of material having electrically alterable characteristics for al- tering the direction of radiation from the antenna structure. 25. The transmission line structure of claim 2 and additionally including a section of anisotropic material selectively located in said load- ing strip layer along its length for inhibiting the propagation of energy along said guiding layer in one direction but not in the other direction, thereby providing a signal isolator structure. 26. The transmission line structure of claim 25 wherein said section of anisotropic material comprises a section of magnetically biased ferrite material. 27. The transmission line structure of claim 26 and additionally including a section of energy absorber material located adjacent one side surface of said section of ferrite material whereby energy propa¬ gation in one direction will be crowded toward said one side surface and be absorbed by said absorber material while energy propagation in the other direction will be crowded toward the other side surface of said section of ferrite and will propagate normally. 28. The transmission line structure of claim 26 wherein said section of ferrite material includes a taper for providing an impedance match between said section of ferrite material and said dielectric loading strip layer. 29. The transmission line structure of claim 28 wherein said section of ferrite material is tapered at both ends. 30. The transmission line structure of claim 26 wherein said section of ferrite material is selectively varied in height to provide additional impedance matching and selected bandwidth characteris¬ tics. 31. The transmission line structure of claim 2 wherein said dielectric loading strip layer is comprised of a first strip of biased anisotropic material and additionally including a second strip of biased anisotropic material having a conductive coating on the top surface thereof located in close proximity to said first strip so as to provide selective signal coupling therebetween as a function of an applied bias to said strips, both said strips terminating at the edges of said guiding layer and said substrate layer, thereby providing a circulator structure. 32. The transmission line structure of claim 31 wherein said first and second strips of biased anisotropic material are comprised of strips of mag¬ netically biased ferrite material whereby asymmetric field distribution produces strong electromagnetic coupling between transmission portions of the guiding layer underlying said first and second strips of ferrite material for signals traveling in but one direction depending upon the direction of magnetiza- tion, whereby selective signal coupling to and from said edges of guiding layer and said substrate layer is provided by control of said direction of magnetiza¬ tion. 33. The transmission line structure of claim 32 and additionally including means for providing a relatively gradual reversal of said magnetization in response to a control signal for providing a signal modulator. 34. The transmission line structure of claim 32 and additionally including means for providing an abrupt reversal of said magnetization in response to a control signal for providing a signal switch. 35. The transmission line structure of claim 32 and additionally including a section of dielectric material between said strips of ferrite material. 36. The transmission line structure of claim 32 and additionally including a section of ferrite material between said strips of ferrite material. 37. The transmission line structure of claim 2 wherein said loading strip layer comprises a strip of light sensitive semiconductor material and wherein said conductive coating on said strip includes a gap along its length located in a light path, said gap and said strip of semiconductor material at said gap receiving light energy in a predetermined mode from an external light source whereby the incidence of light energy at said gap causes a hole-electron plasma to be generated in the semiconductor material to bridge said gap and thereby change its transmission characteristics for the propagating mode in said guid¬ ing layer at the location of said gap, whereby a light controlled device is provided. 38. The transmission line structure of claim 37 wherein said mode of propagation comprises one propagating state in absence of incident light energy at said gap and another propagating state when incident light energy is present at said gap. 39. The transmission line structure of claim 38 wherein said one propagating state comprises the cut-off state and additionally including another loading strip layer and respective conductive coating thereon located adjacent said strip of light sensitive semiconductor material on the input signal side thereof whereby input energy propagating along said guiding layer is reflected at the location of said gap in said cut-off state and coupled to the region of said guiding layer underlying said another loading strip layer. 40. The transmission line structure of claim 39 wherein said another loading strip layer includes a terminal end and additionally including a termination coupled to said terminal end for absorbing energy coupled to said region of said guiding layer underlying said another loading strip layer. 41. The transmission line structure of claim 40 wherein said termination comprises a layer of lossy material located on said guiding layer in contact with said terminal end of said another loading strip layer. 42. The transmission line structure of claim 39 wherein said another loading strip layer comprises a layer of dielectric material. 43. The transmission line structure of claim 37 wherein said propagating mode is modulated. 44. The transmission line structure of claim 2 wherein said guiding layer additionally includes an outwardly projecting energy transition section whereby energy may be coupled to or from said structure. 45. The transmission line structure of claim 44 wherein said transition section comprises a wedge section of predetermined length and having parallel top and bottom surfaces coextensive with said parallel top and bottom surfaces of said guiding layer. 46. The transmission line structure of claim 45 wherein said wedge section further includes a pair of side surfaces intersecting at a predetermined angle for causing input energy impinging on said side surfaces to be totally reflected internally of said wedge section and into said guiding layer. 47. The transmission line structure of claim 2 and additionally including an elongated conductive strip located between said guiding layer and said loading strip layer for coupling energy between said guiding layer and an external device. 48. The transmission line structure of claim 47 wherein said elongated conductive strip has a width equal to or less than the width of said loading strip layer and extends at least to the edge of said guiding layer and substrate layers for connection to said external device. 49. The transmission line structure of claim 48 wherein said elongated conductive strip includes a taper at the inner end thereof for providing an impedance match between said guiding layer and said external device.

Inventors: SEQUEIRA HERMANN BRIAN, SEQUEIRA, HERMANN, BRIAN
Applicants: MARTIN MARIETTA CORP, MARTIN MARIETTA CORPORATION
Application year: 1985
Main classification: H01P3
IPC codes: H01P1, H01P3, H01Q13