A BI-DIRECTIONAL MULTI-FREQUENCY RIPPLE CONTROL SYSTEM

Abstract: A ripple control system for an electric power distribution system wherein the receiver (32) has a pair of synchronous filters (35) driven 90° out of phase with each other, the outputs of which are squared and summed to form an output indepen­ dent of phase, and further discrimination means (44) com­ prising a microcomputer programmed for frequency check­ ing of each synchronous filter output whereby frequency dif­ ferences between the synchronous filter centre frequency and the incoming signal frequency are used to give complete rejection characteristics, thus producing a very narrow bandpass filter for the receiver detection system and enabl­ ing the use of a large number of closely spaced frequencies for the formation of signalling codes.

Description: Tnis invention reLates to improvements in a ripple control system, that is, a means and method for transmitting and receiving information through an alternating current power distribution system. BACKGROUND 0 THS INVENTION Ripple control has been used in the past to transinit information from a power source to a load and thus afford remote control of that load, for example for control of water heaters, street lighting or the like. Reference can be made to our Australian Patent No 531592 (47714/79) (correspondin3 to UK 2022390, USA 435954, Canada 1127710, Switzerland 533333 and otter overseas patents) which descrioed and claimed a system for the control of loads wherein sequential wave form distortion (SWD) signals were transmitted from a power sub-station to a consumer, a transmitter at the sub-station being coupled to the system to derive power from the system and generate a signal the frequency of which was synthesised fro.n and locked to the power frequency, imposing the signal on the power frequency wave form as a sequence of wave form distortions, and the receiver was also coupled to the system and had a detector circuit comprising a pair of correlators (synchronous filters) driven 900 out of phase with each other by a control frequency synthesised from and locked to the power frequency and synchronous with the signal frequency, such that te signal input to the correlators appeared as a D.C. voltage output t@ereof, a squaring circuit was coupled to the signal output of each correlator and a summing circuit coupled to the squaring circuits, arranged so as to square and sum the D.C. voltage outputs of the correlators. The differences between the sequential wave form distortion (SWD) control and conventional ripple control are that the signal was produced as a sequence of wave for distortions by applying a controlled oscillatory load to the power system, the frequency of the signal being locked to the supply frequency1 and the synchronous correlation method of signal detection was used. A series of tests have indicated that the main advantage of the SW3 control over other (prior art) ripple control is its simplicity and lower cost, and also that ost of the receiver circuit can be contained in a custom integrated circuit together with a microcomputer chip and no components are critical. The signal magnitude required for S control is about 10 to 20 percent of that required for ripple control and the reduced signal magnitude avoids :nany of the problems of interference with consumer apparatus sometimes experienced with ripple control. BRIEF SUMMARY OF THE INVENTION The main object of this invention is to provide iaprovements to the SD control system of aforesaid Patent 53592. In that patent there was described a synchronous filter wherein a pair of correlators were in quadrature and thereoy were not phase sensitive. A code was utilised, employing closely spaced frequencies. In this invention, improvements comprise a receiver wherein the receiver has a pair of synchronous filters driven 900 out of phase with each other, the outputs of which are squared and summed to form an output independent of phase, and further discrimination means comprising a microcomputer programmed for frequency checking of each synchronous filter output whereby frequency differences between the synchronous filter centre frequency and the incoming signal frequency are used to give complete rejection characteristics, thus producing a very narrow bandpass filter for the receiver detection system and enabling the use of a large number of closely spaced frequencies for the formation of signalling codes. This improvement results in the synchronous filter of the receivers having a much narrower band pass characteristic. This enhancement allows the full noise rejection advantage of synchronous correlators to be retained together with non-phase sensitivity. The filter is so selective that frequencies as closely spaced as 0.2 Hz can be used to form a multifrequency system of coding. This results in a very large number of usable frequencies between 200 and 1000 Hz and hence groups of unique frequencies can be assigned to each sub-station of a power system and thus overcomes all problems due to spillover of signal from one substation to the next via the high voltage supply system. More specifically, the invention consists of a transmitter for transmitting a code and a receiver for receiving said code, said transmitter being coupled to the distribution system to superimpose a signal on the power frequency waveform as a sequence of waveform distortions, said transmitter embodying frequency control means to accurately generate closely spaced frequencies to enable the use of multifrequency codes, and improvements comprising an input detector circuit in said receiver having two synchronous filters driven ninety degrees out of phase with each other, means for squaring andfsumming the synchronous filter outputs to form a non phase sensitive output, discrimination means comprising frequency checking means for the checking of frequency of each synchronous filter output whereby frequency differences between the synchronous filter centre frequency and the incoming signal frequency are used to give complete rejection characteristics to the filter thus producing a very narrow bandwidth filter. The invention further relates to improvements enhancing the operation of the oscillating load transmitter circuit. By suitable arrangement of the thyristor firing signals the level of oscillation in the transmitter circuit can be controlled. This allows the level of oscillation to be set electronically rather than by heavy current component design. These two enhancements enable the scheme to be used also for the transmission of information upstream from a load (ie consumer) to a power source (ie substation). Because oscillation of the transmitter circuit can be enhanced, circuit resistances are no longer critical to the level of oscillation. This is a significant factor in designing a low cost transmitter. Because multifrequency coding can be used decimal data (eg a meter reading) can be transmitted at a speed which is acceptable. Data pertaining to various other aspects of electrical power transmission can also be transmitted back to a central sub-station, reducing the need for an operator to travel. The transmitter and receiver costs for equipment in accordance with this invention are much lower than for conventional ripple control equipment. The extremely narrow-band pass filter of the receiver not only increases avilability of a large number of signal frequencies but also gives much better rejection of noise signals than normal ripple control receivers and because of this a much lower level of signal power can be used. The low signal power enables a small transmitter to be used and this contributes to the low cost. The two correlators (ie, synchronous filters) can be built onto a silicon chip and require no critical components for tuning thus contributing to the low cost of the receivers, and this is enhanced by a micro-computer algorithm. The transmitter may be of very simple design and consist of a capacitor in series with an inductor and switched to the power supply system with a pair of inverse parallel connected thyristors. Control of the thyristors is such that the circuit is made to oscillate in a discontinuous manner at a selectable signal frequency. Critical tuning of the capacitor and inductor is not required, since the resonant frequency thereof is always greater than the signal frequency. By using different signaling frequencies at each substation the problem of signal spillover from one sub-station to another is removed. It is, however, important that the signal itself should lie within close limits, and, although use can be made of crystal control means to control signal frequency, the control means referred to in said patent 531592 is equally as effective, and more cost effective than crystal control means. The system synthesises identical control frequencies in both the transmitter and receiver. This control frequency is synthesised by multiplying the power supply frequency by an integer using a phase locked loop multiplier circuit and then dividing by another integer. The control frequency is then used in the transmitter to control the oscillation of the inductive capacitive load and in the receiver to control the centre frequency of the synchronous filter. In this way a very precise signal frequency can be transmitted which can be detected by a very narrow band-pass filter. The input element of the receiver may consist of two synchronous filters each consisting of a pair of solid state (eg COOS) change-over switches between a resistor and a capacitor. The change-over switches are driven by the control frequency but one filter is driven 900 out of phase with the other. The output frequency of these two filters will be equal to the difference between the signal frequency and the central frequency. Hence if the signal frequency is identical to the control frequency then the filter output frequency will be zero, that is a D.C. voltage. The total sinal received is formed by combining the outputs from the two filters such that if P and Q are the separate outputs from each synchronous filter and S is the combined output then EMI7.1 The discrimination range of the overall filter is made narrower by checking the frequency of both P and Q by timing between their zero crossings. The signal is rejected if the frequency of both P and Q is higher than a predetermined difference frequency. It is this feature which achieves the very narrow-band pass frequency rate, and in turn makes available a large number of signal frequencies to be usable in a CODE, even though the signal frequency may be below 1000 Hz. Availability of a large number of CODES makes it viable to signal upstrea data from the consumer. When transmitting from a power source, for example a sub-station, the discontinuous oscillatory current in the large transmitter will distort the power system voltage in proportion to the rate of change of current, since the source impedance is almost entirely inductive. Hence a signal of the desired frequency is generated on the power system voltage as a sequence of wave form distortions. A similar system is used for transmission in each direction. There is a difference, however, in that when transmitting "downstream" from a power source to a load the signal appears as a voltage superimposed on the power system voltage and when transmitting "upstream from a load to a power source the signal appears as a current superimposed on the power system load current. When transmitting from a load, that is from a consumer to a sub-station, the discontinuous oscillatory current of the small transmitter is isolated from the high voltage distribution lines at the sub-station by means of a current transformer. The output of the current transformer is passed through an impedance to convert it to a voltage then high pass filtered to remove most of the 50 Hz supply frequency. After amplification it is detected with¯the synchronous filter arrange.sent. BRIEF DESCRIPTION 0? THE DRAWINGS Reference is now made to the accompanying drawings, in which: FIG. 1 shows the basic transmitter, FIG. 2 shows, in simplified form, the system voltage and load current relationship of the basic transmitter, FIG. 3 shows the system arranXement for downstream transmission, FIG. 4 is a block diagram of the receiver, FIG. 5 is a circuit diagram of the receiver power supply, FIG. 6 shows the phase locked loop frequency synthesiser, FIG. 7 shows the synchronous filter arrangement used in the receiver, FIG. 8 shows the synchronous filter output, FIG. 9 shows the arrangement used for signal measurement, FIG. 10 shows the waveform of the two synchronous filter outputs herein designated P and Q, FIG. 11 shows the narrow band pass filter available with the invention, illustrating, by way of comparison, a typical prior art band pass width curve; a band pass width curve obtainable with our earlier Australian Patent 531,592; and in the very narrow band-pass width obtainable with this invention, FIG. 12 shows the current sensing means, FIG. 13 shows the contact sensing means, FIG. 14 indicates the basic arrangement for upstream signalling, FIGS. 15A and 15B show a detail drawing of the overall receiver circuit, FIG. 16 illustrates the relationship between power frequency, thyristor firing pattern frequency, and oscillating current in the LC circuit, FIG. 17 shows the waveform when V1 is high (above) and low (below) (compare FIG. 16), FIG. 18 shows the pattern used for intermediate frequencies (eg 400 to 550 Hz), and FIG. 19 shows the pattern used for higher frequencies (e above 550 Hz). In the basic transmitter 20 (FIGS. 1, 2 and 3), a signal is produced on the distribution bus 21 (11 < V or equivalent) of a sub-station which is then propagated via the 11 kV feeders 22 and 11 000/415 volt distribution transformers 23 to consumers. The signal appears at consumer premises between 240 volt active and neutral on all three phases simultaneously. With SWD control, the signal generating circuit is a switched LC load 24 25 and results in a substantially constant load current I,. The basic transmitter arrangement (in single phase form) is shown in FIG. 1 and the load current relationship to the system voltage in FIG. 2. The transformer load consists of an air cored inductance 24 in series with a capacitor 25 and switched by means of inverse parallel connected thyristors 26. As shown in FIG. 2, the load current does not flow continuously but has an "offs period 27 between each pulse. This off period is achieved by making the resonant frequency of the LC co.bination higher than 1.15 times the control frequency. The actual resonant frequency is not critical and hence it is not necessary to use expensive critically tuned components. The off period ensures the successful operation of the circuit and contributes to the simplicity of the design. Its presence causes the LC circuit to receive energy from the 50 Hz supply and hence oscillation can be maintained. The start of each current pulse is controlled by the firing of one or other of the thyristors by the control frequency (signalling frequency). The thyristors are commutated off as the current pulse goes to zero. This off period is varied with the arrangements which are described hereunder with reference to rIGS. 17, 18 and 19. The operation of this circuit can be enhanced by suitable control of the thyristor firing signals from the transmitter control unit 29. This control fires the thyristors in a pattern, the pattern frequency being the same as the power system frequency V but phase shifted from it by angle ' yes to V' as shown in FIG. 16. The thyristors are fired such that when V' is high oscillation current is continuous from the positive half cycle to the negative, and when V' is low oscillation current is continuous from the negative half cycle to the positive (FIG.17). By varying the phase shift between V and V', ie e, the magnitude of oscillation current in the LC circuit can be varied over at least a 4:1 range. In this way the oscillation current can be set to the desired level. Further variations on this theme are then used depending on signal frequency. The arrangement just described is used for the lower frequencies 200 to 400 Hz. For intermediate frequencies 400 to 550 Hz the arrangement of FIG. 18 is used. In this arrangement continuous oscillation is allowed for each half cycle of the 50 Hz pattern control signal V' except that a full cycle gap follows each transition of V'. In this arrangement the resonant frequency of the LC combination is close to the control frequency. or the higher frequencies, 550 Hz and above, the arrangement of FIG. 19 is used. This arrangement is similar to the first arrangement (ie that used for low frequencies) except that the gap here lasts for a full cycle of the oscillation current and the resonant frequency of the LC combination is close to the control frequency. The control frequencies for firing the thryistors are always derived from the 50 Hz supply and hence are always locked to it. These frequencies are obtained by first multiplying the 50 Hz by an integer in a phase locked loop circuit 30 (FIG. 6) and then dividing by a different inter. In the system herein described a fixed multiplier of 4096 is used and the different frequencies are then formed by dividing by a factor KD. Hence the basic frequency equation is Fc = 50 . 4096 = 204800 .KD KD Eight frequencies are allocated to each sub-station and the same transmitter LC components can be used for each group of 8. Only the control frequency need alter. As the source impedance of the power system (ie from the generators to the 11 kV bus 21) is almost entirely inductive, current I2 in FIG. 1, reflected from the oscillating load, will produce a voltage distortion on the 11 kV bus which is proportional to the rate of change of current Cie VD = Ls dI2) Hence a voltage of the same dt frequency as the current will be produced on the 11 kV bus. In a power system, this arrangement is implemented on a o three phase basis, the current in each phase being 120 apart at the signalling frequency. The system arrangement is shown in FIG. 3. The minimum magnitude of signal voltage required is 0.16% of system voltage with the receivers set at a threshold level of 0.08%. A CODE consists of either 2 or 3 bits, each bit consisting of the transmission of a frequency for a set period of time. A CODE consists of a START bit and either onèor two CODE bits. The duration of each bit is critical. There are 5 different lengths of START bit each defining a different group of functions. There is only 1 length of - CqDE bit. The received duration of all bits must be between specified limits. The spacing between the START bit and the first CODE bit is also used to define different functions. Four different spacings are used. Hence the basic code is as follows: START BIT CODE BITS EMI13.1 <tb> f <SEP> f <tb> <SEP> 1 <SEP> 2 <SEP> 3 <tb> P <SEP> P <SEP> P <SEP> P <tb> <SEP> 1 <SEP> 233 <SEP> <tb> Addresses are formed by defining frequencies f , f and 1 2 f of the code. 3 Functions are formed by defining the periods P and P 1 2 By allocating 8 frequencies to each sub-station the following can be formed with the above code: 512 addresses 20 functions at each address. The 20 functions are arranged as follows: FUNCTION START BIT SPACING NUMBER OF NUMBER OF OPERATE NO. DURATION BEIWEEN CODE BITS ADDRESSES TIME P. - SECS. START & AFFECTED SECS. FIRST CODE BIT P2 SECS 1 5 0 2 1 12.1 2 5 1.2 2 1 13.4 3 5 2 2 1 14.2 4 5 2.8 2 1 5 6.4 0 1 64 10.0 6 6.4 1.2 1 64 11.2 7 6.4 2 1 64 12.0 8 6.4 2.8 1 64 12.8 9 7.8 0 1 8 11.4 10 7.8 1.2 1 8 12.6 11 7.8 2 1 8 13.4 12 7.8 2.8 1 8 14.2 13 9.2 0 1 1 12.8 14 9.2 1.2 1 1 14.0 15 9.2 2 1 1 14.8 16 9.2 2.8 1 1 15.6 17 10.6 0 1 8 14.2 18 10.6 1.2 1 8 15.4 19 10.6 2 1 8 16.2 20 10.6 2.8 1 8 17.0 All Dits' are terminated by reversing the phase of the transmitted signal for 0.6 secs. For example the first START oit listed above consists of EMI15.1 <SEP> f1 <SEP> f1 <SEP> #180 <tb> 4.4s <SEP> 0.6s <tb> This helps to precisely define the duration of each bit, by producing the sharp cut-off shown in FIG. 8, which compares the otherwise long decay time, shown in dotted lines. The total duration of each CODE bit is 3.5 secs ie 3.0 secs followed by 0.5 secs with phase reversed. The basic formula for determining frequency is Fc = 204300 where K is an inter. K D D Frequencies above 200 Hz have ben formed into GROUPS each GROUP consisting of 8 frequencies formed by 8 adjacent values of KD. The frequencies closer than +10 Hz of odd and + 5 Hz of even harmonics of 50 Hz are avoided as they occur naturally on the power system. A binary divider is used in the receivers for KD and the boundaries of each GROUP are determined when the 3 least significant bits of KD are 000 and 111. A 10 bit programmable divider is used for K D and controlled as follows: b9 b8 b7 b6 b5 b4 b3 b2 b1 b0 Frequency X9 X8 X7 X6 X5 X4 X3 X2 X1 X0 f1 as above Y2 Y1 YO f2 as above Z2 Z1 Z0 f3 X, Y & Z are manually programmed into each receiver to set its address code. The selection of which GROUP of frequencies to use for any one sub-station will be determined by the power system characteristics. The main parameter which will affect selection will be the existence of 11 kV capacitor banks on the system, either as a large bank mounted in a sub-station, or smaller banks installed out along feeders. The existence of these banks will make it more economical to use the lower frequencies whereas if capacitor banks are not present, the higher frequencies will result in smaller transmitter components. Capacitor banks installed on the higher system voltage, ie 66 kV, will have negligible effect on signal generation. Since capacitor banks are not part of the invention, they are not illustrated herein. The receiver 32 (FIG. 15) may be divided into 8 separate functional blocks as follows: 1. Power supply 33 (FIGS. 4, 5 and 15). 2. Frequency synthesiser 34 (FIG.15). 3. Synchronous filter 35 (rIGS. 7 and 15). 4. Signal measurement 36 (FIGS. 9 and 15). 5. Computation and logic 37 (FIG. 15). 6. Current sense 38 (FIG.15). 7. Contact sense 39 (FIG.15). 8. Output bi-stable contactor 43 (FIG.13). By using CMOS integrated circuits throughout, the power supply only needs to provide a few milliamps. It can therefore be made very simple by using a capacitive divider, rectifier and regulator direct from the 240V AC supply (see FIG.5). The frequency synthesiser section is built up using a standard phase locked loop multiplier 40 followed by a programmable divider 41 to form the required filter frequency. The 50 Hz is multiplied by 16384 and then divided by an integer KD which is manually programmed in. A This results in a frequency four times the desired filter frequency and is subsequently divided to produce two control signals at the desired frequency but in quadrature (ie one displaced 900 from the other). The selection of frequency Fc1, Fc2 or Fc3 is controlled by the data selectors D3 and D4 (FIG.6). A single stage RC filter and Schmitt trigger circuit 42 are used to condition the 50 Hz input signal from the supply to the phase locked loop (FIG. 6). The low pass filter from the phase comparator output on the phase locked loop to the voltage controlled oscillator (VCO) input needs to have a fairly long time constant for stability. For the same reason the VCO range selection resistors need to confine the VCO frequency to about +10 of centre frequency. The synchronous filters 35 (or correlators) form the heart of the receiver circuit. These circuits extract the signal frequency from the 50 Hz supply and do so in a very selective manner. Each synchronous filter consists of a pair of solid state chanOe-over switches in the middle of an RC filter (Ri C2), followed by a second stage pf RC filtering (R2 C2) for smoothing as shown in FIG. 7. If the frequency driving the switches is the same as the transmitted frequency and-both are in phase then the switches will act as a synchronous rectifier and the rectified signal voltage will appear as a DC voltage on capacitor C2. When the signal frequency is 900 out of phase with the correlator frequency, zero signal voltage will appear on C2. If the correlator frequency is very slightly different from the transmitted frequency then, in effect, their phase relationship will change through 3500 at a rate equal to the difference frequency, with the result that the voltage appearing on capacitor C2, will be low frequency AC. By using two synchronous filters driven by control 0 frequencies which are in quadrature (90 apart) and taking the square root of the sum of their squares, a non-phase sensitive detector is formed. EMI18.1 where S = total signal P and Q are outputs from each synchronous filter. By time checking the zero crossing of each correlator output (ie P & Q) separately, a limit can be set on the low frequency AC output referred to above. In this way the filter characteristic can be made extremely narrow and complete rejection of all frequencies outside the pass band can be obtained (FIG.11). For example, with a time check of 2.4 seconds, complete rejection of all frequencies outside of +0.2 Hz of the centre frequency is obtained. As the 240 V 50 Hz voltage is very much larger than the signal voltage it will appear on capacitor C1 as a ripple voltage. A second stage of RC filtering (R2 C2) is used to eliminate this ripple. The total synchronous filter circuit, as shown in FIG. 7, gives ground referenced outputs at P & Q which are then sampled into the measurement circuit. Due to the RC time constants of the synchronous filters, outputs P and Q will have slow rise and fall times. o By reversing the phase (ie 180 phase change) of the transmitted signal for the last 0.6 secs of each bit the fall times of P and Q can be made much shorter and the signal duration above threshold then becomes almost independent of signal magnitude. This mininises transmission time and allows bit duration to convey information. This is referred to above and illustrated in FIG. 3. The signals P and Q from each of the synchronous filters are measured alternately each second cycle of the 50 Hz, ie every 40 ms. A digital to analogue converter circuit is used controlled by a successive approximation routine in a microcomputer. FIG. 9 shows the circuit used. All the computation and logic for the receiver is carried out in the microcomputer 44. (Microcomputer 44 is a Philips PCD3343, available from Philips, Eindhoven, Holland). As mentioned above the microcomputer 44 takes a digital measurement of each of the two synchronous filter outputs. The total signal S is then computed from EMI20.1 In addition, when S is below threshold long running averages, using 256 samples, are also computed of P and Q and these values then used for zero correction. For most of the tine no signal is present and hence if the signal is averaged with a long time constant it represents the residual signal. Such a residual can be present due to cross talk and leak current in the switches, bias current in the comparator and offset voltages in the comparator and operational amplifier. If the averaging time constant is made very long compared to signal duration the running average will not change appreciably during the receipt of a signal. The residual measurements for P and Q are then subtracted from P & Q to give corrected readings. The signal is tested for magnitude and the time it is above a set threshold is measured. The CODE bits have the shortest duration and they require that the signal is above threshold for at least 2.4 seconds but not more than 3.8 seconds. START bits with five different durations are used and they have the following limits. START BIT MIN. DURATION SECS. 1 3.8 2 5.2 3 6.6 4 8.0 5 9.4 The improved frequency discrimination i/s achieved by checking the frequency of each of the correlator outputs, ie P & Q. As explained before if the incoming signal frequency is slightly different from the synchronous filter frequency, then P and Q outputs will be at the difference frequency. This frequency can be checked by using the P and Q zero crossings to logically develop a rejection signal R as shown in FIG. 10. This logic is performed by a routine in the microcomputer and uses three flags, FT the threshold flag, FP the P flag and FQ the Q flag. If total signal S is below threshold all three flags are reset and rejection signal R is pulsed high. When S is above threshold flags FP and FQ are set respectively whenever P or Q perform a zero crossing. If when FP is set FQ is already set then R is pulsed high and FQ is reset. Similar conditions apply if FP is already set when FQ is set. The pulsed signals R are used to reset a signal check timer thus rejecting the signal. Therefore in order for a signal to be accepted the time between reset pulses must be longer than the required bit time. For a CODE bit the maximum check time is 3.8 seconds which corr-sponds to a period for P and @ of 2 Ä 3.3 = 7.6 secs which represents a frequency of O.132 Hz. Tha is if the incoming signal is further removed than +0.132 Hz from the synchronous filter frequency it will be rejected. Similarly for the longest START bit the maximum check time is 10, seconds and in this case if the incoming signal is further removed than + 0.04s Hz from the synchronous filter frequency it will be rejected. The above frequency checks on P and Q result in the filter characteristic shown in FIG. 11. In FIG. 11, the broad band pass width of prior art is graphically depicted. This was greatly narrowed in our earlier development the subject of Australian Patent 531592, and is further narrowed in this invention as shown in FIG.11 As eacn bit is received the microcomputer decides whether it is a START or a CODE bit according to its duration. All decoding of the signal is done by the microcomputer 44. The 50 Hz going in to the phase locked loop from the 16334 divider is compared in phase with the 50 Hz supply input. If the phase difference is greater than 3OOs the loop is considered to be OUT OF LOCK. Whenever an out of lock occurs the timers measuring signal duration are reset. When a receiver is used to control a storage hot water system, it is an advantage fro a power system point of view to allow only one heating period during any prescribed ON time. For instance if an ON time is allowed between 10 pm and 7 am, then most heaters will attain full temperature say by 5 am and their thermostats will open. If a household then uses hbt water at say 6 am, the thermostat will close and a second heating period will commence. This second heating period places load on the power system at a time when other loads are heavy (industry just starting up) and should therefore be avoided. The prescribed ON time cannot be made shorter because a small number of consumers will require the full time for their storage system to reach full temperature. To prevent the second heating period a lockout routine is included which senses the condition that no load current is flowing when the contactor is closed (hence the thermostat must be open). Under these conditions the microcomputer operates the contactor to the OFF position. The contact sense and current sense circuits are described later. An over-ride timer routine is included to provide a fail-safe feature in the event of loss of communication with a sub-station. This can occur due to temporary feeder changes or a transmitter failure. This timer causes the output contactor to close 6.2 hours after the receipt of the last OFF command. If the contactor is required to remain off, a signal must be transmitted before the 6.2 hours expires in order to reset the timer. The only signal required to reset the timer is a START bit. A software routine is included in the microcomputer 44 which enables the switch ON of low usage consumers to be delayed. By timing each day the duration the water eater is energized and averaging this over a wee a consumer's usage can be assessed. After an ON signal is sent, actual switch on can then be delayed in inverse proportion to normal usage. All these functions are embodied in the microcomputer software, the program of which forms no part of this invention. This allows the load of low usage consumers to be transferred to the early hours of the morning where there is a "trough" in the daily load curve. Current flow in the load circuit is sensed by means of a small current transformer as shown in FIG. 12. When load current flows output I goes high on a 50 Hz basis. For no load current, I is low. The transition of I to the high state initiates an interrupt routine in the microcomputer. Current sense is used in the "lockout" and "delay" routines. The position (open or closed) of the output contactor 43 is sensed with the circuit shown in FIG. 13. When trite contact is open output C is lo. When the contact i3 closed C gores high on a 50 Hz basis which is sensed by the microcomputer. Contact sense is used in the "decodin" and "lockout" routines. The output contactor needs to be mechanically bi-stable and of the impulse type. That is, it should have only one coil which is energised for a short time (approximately 43 ms) to change the state of the contract. The relay should be as efficient as possible to minimise the power required to drive it. A single make contact is required having a current rating of at least 30 amperes AC resistive. The same principles outlined so far, and method of coding can be used for signaling from a consumer back to a sub-station. This is referred to as upstream signalling. There is one important difference, however, between upstream and downstream signalling. Whereas downstream is a voltage scheme, upstream uses current. In downstream signaling a signal voltage of a very precise frequency is superimposed on the normal system voltage and this is detected by a synchronous filter in the receiver. In upstream signaling a very small transmitter 45, but still of the same configuration as described above, is located at the consumer. The transmitter current is at a very precise frequency and may be detected at the sub-station by first turning it into a voltage with current transformer 47 and then usin the synchronous filter to detect it. The basic arrangement is shown in FIG. 14. As before, a ;nulti-frequency scheme can be use for coding. However if variable data is required to be transmitted rather than control signals to specific addresses then a bank of fixed synchronous filters will be required, one for each frequency used. For example, if 10 frequencies are used to transmit deci.nal information, 10 synchronous filters will be required and each frequency will represent a decimal number. As many different frequencies are available for use, simultaneous transmissions are possible on the same feeder. Simultaneous transmissions usin the same frequencies are also possible on different feeders as each feeder current is entirely separate. It is quite important to note that the thyristor control is such that firing of te thyristors is in a pattern CIGS. t8 and 19), the pattern frequency being the same as the power system frequency but phase shifted from it, suc that for one-half the period of the pattern, the transmitter load current is continuous from the positive half cycle to the negative half cycle and for the other half period transaitter load current is continuous from the negative half cycle to the positive half cycle, the basic firing frequency being the signal frequency required and the resonant frequency of the inductance capacitance combination being about 20% (between 5% and 50%) higher than the signal frequency (for the lower frequencies) thus causing discontinuous oscillation because of the short periods of zera load current. For higher frequencies, the capacitance inductance combination has about the same resonant frequency as the signal frequency and the firming pattern of the thyristors is such that oscillation is continuous except for one complete period of the oscillation following each transition of the pattern frequency. For still higher frequencies the capacitance inductance combination has about the same resonant frequency as the signal frequency and oscillation is only allowed on every alternate cycle. In all instances it will oe noted tat the downstream signals are voltage signals and the @pstream signals are current signals, this being necessary because of tiae inherent impedances of a power supply network. While the consumer receiver and transmitter is separately described above, it will be recognised oy those skilled in the art that a common frequency synthesiser and microcomputer combination can be used both for receiving and transmitting modes. Reversals of phase in the transmitted signal which is referred to above is achieved merely by delaying the firing o of the thyristors by 130

Claims: 1. In a ripple control system for an alternating current electric power distribution system which comprises a transmitter (20) for transmitting a code and a receiver (32) for receiving said code, said transmitter being coupled to the distribution system to superimpose a signal on the power frequency waveform as a sequence of waveform distortion, characterised by the following improvements: : frequency control means (29) to accurately generate closely spaced frequencies in the transmitter to enable the use of multifrequency codes, an input detector circuit in said receiver having two synchronous filters (35) driven ninety degrees out of phase with each other, means (44) for squaring and summing the synchronous filter outputs to form a non-phase sensitive output, discrimination means (44) comprising frequency checking means for the checking of frequency of each synchronous filter output whereby freouency differences between the synchronous filter centre frequency and the incoming signal frequency are used to give complete rejection characteristics to the filter thus producing a very narrow bandwidth filter. 2. Improvements in a ripple control system according to claim I further characterised in that said discrimination means comprises a microcomputer (44) programmed to logically examine the zero crossings of the two synchronous filter outputs to produce rejection of incoming signals when the period between said zero crossings is less than a fixed code bit time. 3. Improvements in a ripple control system according to claim 1 or claim 2 further characterised by a transmitter transformer (28), said transmitter (20) being coupled by that transformer to the distribution system to derive power therefrom and generate signals by means of sequences of waveform distortions, the frequencies of said signals thereby being synthesised from and locked to the power frequency. 4. Improvements in a ripple control system according to claim 3 further characterised in that said transmitter comprises an inductance (24) in series with a capacitance (25), a pair of thyristors (26)connected in inverse parallel and fired by a transmitter control unit (29) in a pattern, said transmitter transformer-(28) coupling the signals generated by said firing to the power system. 5. Improvements in a ripple control system according to a claim 4 further characterised in that said inductance capacitance combination (24, 25) has a resonant frequency higher than said signal frequency and said transmitter control unit (29) is arranged to fire the thyristors at the signal frequency resulting in gaps of zero current following each half cycle of oscillation current. 6. Improvements in a ripple control system according to claim 4 further characterised in that said inductance capacitance combination (24, 25) has a resonant frequency higher than said signal frequency and said transmitter control unit (29) is arranged to fire the thyristors at the signal frequency but additionally controlled such that wtien a logic signal of mains frequency. but phase shifted from the mains, is logic HIGH then oscillation current is continuous from the positive half cycle to the negative half cycle with a gap of zero current following the negative half cycle, and when such logic signal is LOW oscillation current is continuous from the negative half cycle to the positive half cycle with a gap of zero current following the negative half cycle. 7. Improvements in a ripple control system according to claim 4 wherein said inductance capacitance' combination (24, 25) has a resonant frequency near said signal frequency and said transmitter control unit (29) is arranged to fire the thyristors at the signal frequency, but said control unit additionally controls firing of the thyristors (26), such that when a logic signal of mains frequency, but phase shifted from the mains, is logic HIGH then oscillation current is continuous until the logic signal changes to LOW whereupon following the first positive half cycle of oscillation current a gap of zero current ensues for one full signal frequency period, and when said -logic signal changes from LOW to HIGH then following the first negative half cycle a gap of zero current ensues for one full signal frequency period. 8. Improvements in a ripple oontrol system according to claim 4 further characterised in that said inductance capacitance combination (24, 25) has a resonant frequency near said signal frequency and said transmitter control unit (29) fires the thyristors (26) basically at the signal frequency but additionally controls firing of the thyristors, such that when a logic signal of mains frequency, but phase shifted from the mains, is logic HIGH then oscillation current is continous from the positive half cycle to the negative half cycle with a gap of zero current for one full signal period following the negative half cycle and when the logic signal is LOW oscillation current is continuous from the negative half cycle to the positive half cycle with a gap of zero current for one full signal period following the positive half cycle, with an extra oscillation half cycle immediately following each logic transition. 9. Improvements in a ripple control system according to any preceding claim further characterised in that said code comprises a START bit consisting of a signal frequency being transmitted for a set period of time followed by a set period of no transmission followed by the transmission of at least one code CODE bit comprising the tranmission of a second frequency for a set period of time. 10. Improvements in a ripple control system according to claim 7 further characterised in that said code is indentified by the START bit duration and frequency, period between START bit and the first CODE bit, and each CODE bit duration and frequency. 11. Improvements in a ripple control system according to either claim 7 or claim 8 further characterised by phase reversal means (29) by which each said bit is terminated by reversing the phase of the transmitted signal for sufficient time to achieve a sharp cut-off in the received signal. 12. Improvements in a ripple control system according to any one of the preceding claims further characterised in that said frequency control means to control the synchronous filters in the receivers and the thyristor firing in the transmitter control unit is synthesised from the power mains frequency of the electric distribution system firstly by fixed frequency multiplication using a phase locked loop frequency multiplier circuit (40) followed by a variable divider (41) to produce a plurality of control frequencies. 13. Improvements in a ripple control system according to any one of the preceding claims further characterised in that said transmitter (20) is at the power supply end of said distribution system and the signal transmitted is transmitted by said distribution system to a consumer as a voltage signal. 14. Improvements in a ripple control system according to any one of claims 1 to 12 further characterised in that said transmitter is at a consumer end of said distribution system and the signal transmitted is transmitted by said distribution system to the power supply end thereof as a current signal. 15. Improvements in a ripple control system according to claim 14 further characterised by a current transformer (47) at said power supply end transforming said current signal to a voltage signal which is directed into said receiver. 16. A ripple control system further characterised in that it embodies the improvements of any one of claims 1 to 15. 17. A method of transmitting and receiving a ripple control system signal further characterised in that it includes discriminating the transmitted signal at the receiver end by discrimination means (44) according to any one of the preceding claims. 18. A method of transmitting and receiving a ripple control system signal further characterised in that it includes transmitting said signal by means according to any one of claims 6, X, 8, 10 or 11.

Inventors: FOORD, PETER MAXWELL
Applicants: THE ELECTRICITY TRUST OF SOUTH AUSTRALIA
Application year: 1985
Main classification: H02J13
IPC codes: H02J13