RESISTIVE LOOP ANGULAR FILTER

Abstract: An angular filter (50) for electromagnetic radiation is formed of a set of elements supported on a dielectric substrate. The elements are rods (100) or loops which are electrically conducting and include resistance for dissipating energy of the radiation. Each rod (100) is formed as a linear element parallel to the axis (300) of propagation of the radiation. Each loop element is formed as a closed loop in a plane normal to an axis of propagation of the radiation. This minimizes interaction with a transverse magnetic field of the radiation at zero angle of incidence to the filter, the interaction and consequent attenuation increasing with increasing angle of incidence. Thereby, spurious sidelobes of a radiation pattern associated with a radar or other antenna can be reduced by the filter in favor of the main lobe along the antenna axis. The loop elements may also be formed by a set of members spaced apart to introduce capacitance for resonating with inherent inductance of the members, there­ by to enhance the filter attenuation.

Description: This invention relates to the propagation of electromagnetic waves and, more particularly, to an angular filter comprising an array of elements which interact with the electromagnetic waves as a function of the angle of incidence of a wave upon a surface of the filter. An angular filter, also referred to as a spatial filter, is a device which passes or attenuates an electromagnetic wave depending on the angle of incidence of the wave relative to a surface of the filter. Typically, such filters are designed to pass a wave propagating at normal incidence (broadside) and to provide attenuation or rejection that increase with increasing angle of incidence away from broadside. The filter may be employed in combination with a directive antenna of electromagnetic radiation, in which application the filter serves to reduce sidelobes in the radiation pattern of the antenna. Several types of angular filters have been described in the literature including, by way of example, multilayered dielectric filters (R. J. Mailloux, "Synthesis of Spatial Filters with Chebyshev Characteristics", IEEE Trans. Antennas and Progagation, pp. 174-181; March 1976), perforated metal sheet filters (E. L Rope, G. Tricoles, "An Angle Filter Containing Three Periodically Perforated Metallic Layers", IEEE AP-S Int. Symp. Digest, pp. 818-820; 1979) and multilayered metal-grid filters (R. J. Mailloux, "Studies of Metallic Grid Spatial Filters", IEEE Int. Symp. Digest, p. 551, 1977; P. R. Franchi, R. J. Mailloux, "Theoretical and Experimental Study of Metal Grid Angular Filters for Sidelobe Suppression", IEEE Trans. Antennas and Propagation, pp. 445-450, May 1983; P. W. Hannan and J. R. Pedersen, "Investigation of Metal Grid Angular Filters", Proc. 1980 Antenna Applications Symposium, Allerton Park, Illinois, September 1980; and J. F. Pedersen, P. W. Hannan, "A Metal Grid 5 x 5 Foot Angular Filter", IEEE AP-S Symp. Digest, pp. 471-474, 1982). Various forms of construction have been utilized in the fabrication of the angular filters resulting in a variety of benefits and limitations. By way of example, metal-grid angular filters are practical and can offer improved performance, such as a reduction in wide-angle sidelobes, when combined with an antenna. However, the metal-grid filters are limited in the useful frequency bandwidth due to the dependency of the filter characteristics on frequency. Also, such filters have an inherent resonant nature necessitating tight dimensional tolerances in their construction. An insufficiency in the tolerances may result in variations of transmission phase across the filter aperture for angles of incidence within the filter angular passband. Such phase variations can create unwanted sidelobes in the radiation pattern produced by the combination of the antenna with the filter. A further limitation found in filters having the metal grid construction is the rejection of electromagnetic power by reflection rather than by absorption. Such reflected power can return to the antenna, associated with the filter, and then reflect back to the filter. Such multiple reflection yields unwanted sidelobes within the angular passband of the filter. Thus, it is seen that the present forms of construction introduce limitations which detract from the benefits which would otherwise be provided by the angular filters. US-A-4343002 describes a filter for filtering an electromagnetic wave energy signal incident thereon, said filter comprising: a substantially planar array of resistive elements forming a substantially non-reflective array and being capable of interacting with an incident electromagnetic wave; and support means formed of dielectric material, said support means supporting said elements in said substantially planar array and being substantially transparent to an incident electromagnetic wave. Those elements are described as discs which may be fabricated of a material which reflects the radiation, a material which absorbs the radiation, or a dielectric material of a certain stipulated thickness which changes the phase of the radiation by 90°. By this technique, ripples are substantially removed from the beam pattern. This smoothing out of ripples by reflecting or absorbing or phase shifting the incident electromagnetic energy is regardless of the angle of incidence of the wave. The present invention is characterized in that: said resistive elements are adapted such that an electromagnetic wave incident on said filter in a direction normal to said planar array passes through said filter and said resistive elements are adapted to each dissipatively attenuate an electromagnetic wave incident on said filter at an angle to said normal direction. The present invention is applicable to angular filtering for E-plane incidence and for H-plane incidence. These terms are explained below in relation to Figures 2 and 13. The aforementioned aspects and other features of the invention are explained in the following description, taken in connection with the accompanying drawing wherein: Figure 1 is a partial view, in perspective, of an axial conductance angular filter according to the invention. Figure 2 illustrates an electromagnetic wave incident on an angular filter in the E plane of incidence. Figure 3 is a graph illustrating the computed attenuation normalized as to wavelength versus angle of incidence (in degrees) for a homogeneous filter medium according to the invention. Figure 4 is a graph comparing the measured and computed attenuation versus angle of incidence at 5 GHz for a 5 x 5 foot (1.52m x 1.52m) angular filter medium according to the invention. Figure 5 is a graph comparing the measured and computed attenuation versus angle of incidence at 10 GHz for a 5 x 5 foot (1.52m x 1.52m) filter medium according to the invention. Figure 6 is a graph comparing the measured and computed attenuation versus angle of incidence at 20 GHz of a 5 x 5 foot (1.52m x 1.52m) filter medium according to the invention. Figure 7 is a perspective view of a perferred embodiment of a filter medium according to the invention. Figure 8 is a cross sectional view of the medium of Figure 7 taken along lines 8-8. Figure 9 illustrates in partial perspective view the strip-type medium which may be imbedded in a dielectric in accordance with the invention. Figure 10 is a graph illustrating the normalized attenuation versus incidence angle for various values of the axial loss tangent (D). Figure 11 is a stylized view of a radar antenna combined with an angular filter incorporating the invention for the attenuation of sidelobes while permitting the radiation to pass along the main lobe; Figure 12 is an enlarged fragmentary view of a portion of the filter of Figure 11, a part of the view of Figure 12 being cut away to disclose filter elements on different ones of a plurality of lamina of the angular filter; Figure 13 is a fragmentary sectional view of a filter element taken along the line 13-13 in Figure 12; Figure 14 is a plan view of a portion of the surface of the filter of Figure 11 showing the relative positions of a group of circularly shaped radiating elements; Figure 15 shows a plan view of a set of square shaped radiating elements; Figure 16 shows a view similar to that of Figure 14, but presenting a set of filter elements having diameters much reduced from the spacing between elements as compared to the arrangement of Figure 14; Figure 17 shows a form of element being constructed of spaced apart members on both sides of a dielectric layer to provide for capacitance; Figure 18 is a fragmentary sectional view taken along the line 8-8 in Figure 17 showing a gap between two of the arcuate members of the filter element; Figures 19 and 20 show schematically the configurations of two loop elements having both resistance and shielding, there being shielding members external to the loop in Figure 19, the shield being a shorting member in Figure 20; Figure 21 shows schematically the presence of both a capacitive element and a resistive element in a filter element; Figure 22 shows schematically a loop embodying the features of both Figures 19 and 21; and Figure 23 shows schematically a loop having a shorting shielding member and two capacitive elements disposed on each half of the loop. Figure 1 describes an axial conductance angular filter according to the invention. Specifically, an array of axially oriented resistive elements 100 (such as rods or strips) having a certain value of conductance or resistance in the axial direction is embedded in a dielectric supporting material 200. These thin axial elements 100 are neither good reflectors nor good conductors, but rather, provide a certain amount of conductance or resistance in the axial direction. The amount will be described below in detail. A wave 300 at normal incidence (i.e. in the axial direction) does not induce current in the axial resistive elements, and the filter is essentially invisible to this wave. For oblique angles of incidence in the E plane, current is induced in the resistive elements 100 and dissipative attenuation occurs. The angular filter 50 operates over a wide frequency band and does not require tight dimensional tolerances because the dissipative attenuation does not rely on resonance. As indicated in Figure 2, an electromagnetic wave incident on filter 50 in the E plane of incidence has an axial component of electric field which is proportional to sin T, where T is the angle of incidence away from broadside 300. If we assume that this is also true within the filter medium, then the axial current I in the filter should also be proportional to sin T. Since this current flows through resistive elements, there is power dissipated within the filter. This dissipated power should be proportional to I² and hence proporational to sin²T. This heuristic analysis neglects to account for the effect of the axial-conductance medium on the incident wave, and it does not relate the dissipated power to the incident power. Nevertheless, the sin²T proportionality is a fairly good approximation for the dissipative loss of the axial-conductance angular filter 50. Assuming that the sin²T proportionality represents the dissipative loss of an axial-conductance filter, we can expect that filter 50 should provide continuously increasing rejection with incidence angle in the E plane. This desirable result does not always occur with other types of angular filters. For example, the multilayer dielectric filter is subject to Brewster-angle effects in the E plane of incidence, and the crossed metal-grid filter may provide little or no rejection near grazing incidence in the E plane. Another feature that can be anticipated for axial-conductance filter 50 is that it should be inherently invisible at broadside incidence. This is a result of its thin axially-oriented elements which have essentially no effect when the electric field is perpendicular to them. Such a filter, when placed in the aperture of a narrow-beam antenna, should have only a small risk of adversely effecting the main beam or raising the nearby sidelobes. A corollary of this inherent broadside invisibility is that axial-conductance filter 50 does not have critical tolerances on dimensions or materials. Variations of filter thickness or resistance values do not affect the amplitude or phase of the main-beam power passing through the filter near broadside incidence, so no new sidelobes are created. Only the wide-angle rejection value would be affected, which is not a critical factor. Still another feature that can be anticipated for axial conductance filter 50 is that its rejection of incident power will occur primarily by means of absorption. Reflection from the filter for most angles of incidence will tend to be fairly small. This reduces the chance that rejected power will return to the antenna and then be re-reflected to create new sidelobes. Finally, it can be anticipated that axial-conductance filter 50 would provide all of the above features over a wide frequency band. Since its operation does not depend on a resonance or a grating-lobe phenomenon, it is not strongly affected by a change of frequency. There is a certain relation between wide-angle rejection and frequency, but this can still permit a wide useful frequency band of operation. The features mentioned in the previous paragraphs involve some limitations that do not occur with other types of angular filters. One limitation of axial-conductance filter 50 is that it provides rejection versus angle only in the E plane of incidence. Another limitation is that a sharp increase of rejection with incidence angle (i.e., a sharp cutoff) is not obtainable, unless some resonant or frequency-sensitive mechanism is incorporated into the filter medium. Even with these limitations, the positive features of axial-conductance filter 50 make it worthy of consideration for use either alone or in combination with another filter. Each resistive element 100 should have a substantially low conductivity. In particular, the range of the conductivity of the resistive elements can be defined as follows. If the dielectric 200 is assumed to have an effective permittivity approximately equal to that of free space and the resistive elements 100 embedded therein are assumed to form a filter medium which is homogeneous with a certain axial conductance (S ax ), the attenuation constant (A) in the medium (in napiers per meter) can be derived as a function of the E-plane incidence angle (T): Where W is the frequency of the incident electromagnetic energy in radians per second and E o is the permittivity (or electric constant) of free space and λ is the wavelength of the incident wave in meters. The parameter S ax /WE o is the axial loss tangent (D) of the medium. Figure 3 is a graph illustrating computed curves of attenuation in decibels per wavelength of filter thickness versus T for various values of the axial loss tangent (D). It can be seen that a value for D near unity is preferred and that the actual value of D is non-critical and may be in the range of 0.5 to 2.0 while yielding nearly optimum performance. A comparison of the several curves in Figure 3 at small incidence angles confirms that D = 1 gives the greatest attenuation at small angles. Also, the D = 1 case gives almost, but not quite, the greatest attenuation near 90 o incidence. The curves of Figure 3 give essentially the angular rejection characteristic of a filter using an axial-conductance medium. For example, with a medium having D = 1, a rejection of almost 8 dB would be obtained for a wavelength-thick filter at 45 o incidence. For a filter two wavelengths thick, almost 16 dB would be obtained at 45 o . At 90 o , the attenuation for the D = 1 case is about twice the value at 45 o . In addition, there would be a substantial reflection loss near 90 o . There is no indication in any of the curves of Figure 3 that the filter rejection might decrease with increasing angle (as it can with some other types of angular filter). Near 0 o incidence, the filter attenuation characteristic is inherently square-law with angle. For a filter two wavelengths thick, the attenuation of the homogeneous axial-conductance medium would be less than 0.1 dB over a ± 3 o range of incidence angles centered on broadside. Thus a pencil-beam antenna having a beamwidth of 3 o or less should have virtually no change of peak gain when operated with such a filter over its aperture. The shape of the curves in Figure 3 is of some interest. To compare the shapes for different values of D, the attenuation of each curve can be normalized to its value at 90 o incidence. Figure 10 shows the resulting set of curves. Also shown is a sin² T curve. It is evident that for values of D equal to unity or more, the sin² T curve gives a good approximation to the actual shape of the A versus T curve. The approximation becomes poor for values of D much less than unity. Another question is: how does the rejection at some angle vary over a wide frequency band? The answer to this question is contained in the curves of Figure 3. It is evident that the basic factor is attenuation per wavelength of the medium. Thus, for a filter having a specified thickness (in inches), the principal term is a linear increase of attenuation with frequency. A secondary term also exists because D is inversely proportional to frequency. However, if D is set to unity at midband, the variation of D that would occur over a frequency band as much as two octaves wide would still have only a relatively small effect on attenuation. This is another case in which the non-critical nature of D is helpful. The actual inhomogeneous medium illustrated in Figure 1 is more difficult to analyze and its performance is more complex. However, when the resistive elements 100 are thin and are closely spaced relative to the wave length of the incident electromagnetic energy, the performance approximates that of the homogeneous medium as given in Figure 3. Dielectric material having an effective permittivity substantially greater than that of free space also modifies the performance. In order to understand the relationship between elements 100 and the axial loss tangent (D), it is helpful to define a quantity R λ as the resistance (in ohms) across a cube having wavelength sides. The quantity R λ is equal to the axial resistivity divided by wavelength, and hence equals l/S ax λ. Defining the axial loss tangent (D) as equal to S ax /WE o , the relation between R λ and D is then obtained: If a value of unity for D is wanted, then the medium should provide a resistance of 60 ohms in the axial direction between opposite faces of a wavelength cube. The resistance elements can have any convenient cross-sectional shape. In a preferred embodiment thin strips are selected because such strips can be produced by printed-circuit techniques. Figure 9 is a partial perspective drawing showing an array of resistance strips comprising the inhomogeneous axial-conductance medium. The array lattice is square with spacing s, and the width of each strip is w. It is assumed that the strips are very thin, and that their resistance behavior can be defined in terms of the surface resistance R s (in ohms per square) of the strip material. The following relation can then be derived: R λ = (s/λ)² λ w R s (2) Combining (1) and (2) yields a formula for R s in terms of D and the array/strip dimensions: As an example, suppose that s/λ = 0.2, and w/s = 0.2, and a value of unity for D is wanted. Equation (3) then yields 60 ohms per square as the surface resistance needed for the strip material. A filter 5 feet by 5 feet (1.52m x 1.52m) in aperture size and 5 inches (12.7cm) in thickness was developed for operation at 10 GHz. Resistive elements 100 of the developed filter were screen printed on thin dielectric sheets which were stacked alternately with foam spacers as shown in figures 7 and 8. In particular, thin dielectric sheets 201 were screen printed so that resistive elements 101 were located on one surface thereof. Stacked between successive sheets 201 were dielectric sheets of foam spacers 202. This assembly was enclosed within a protective fiberglass shell and contained over 70,000 printed resistive elements 101. The attenuation of the constructed filter was measured versus E-plane incidence angles at 5, 10 and 20 GHz. Figures 4, 5 and 6 show the measured attenuation points together with curves computed from the homogeneous medium analysis. Reasonable similarity between the two is evident. Additional measurements of filter samples in simulator wave-guide have yielded results similar to the computed values out to angles close to grazing incidence, where the panel measurements are difficult to obtain with accuracy. Thus, the axial conductance angular filter according to the invention has a yielded satisfactory and useful angular rejection characteristic over a two-octave bandwidth. The angular filter according to the above embodiment of the invention has been generally described as an array of parallel resistive elements 100 supported in dielectric material 200 being parallel to the normal of the sheet. The invention contemplates that more than one array of parallel resistive elements may be embedded in the dielectric and that the orientation of the resistive elements does not necessarily have to coincide with the direction perpendicular to the face of the dielectric. Figure 11 shows a radar antenna 20 having a dish 22 which serves as a radiating aperture for radiating a beam 24 of radiation. The beam 24 is characterized by a main lobe 26 and sidelobes 28. An angular filter 30 incorporating the invention is positioned in front of the dish 22 and carried by the antenna 20 for improvement of the shape of the radiation pattern of the beam 24. In Figure 11, the antenna 20 and the filter 30 are shown in exploded view so as to disclose a front surface 32 of the filter 30. In accordance with the invention, the filter 30 comprises a set of laminae 34 of dielectric material which is transparent to the radiation of the beam 24, the laminae 34 being arranged serially along an axis 36 of the dish 22 with their surfaces parallel to the front surface 32 and normal to the axis 36. Each lamina 34 supports an array of filter elements 38 which interact with the magnetic field vector H but with minimum interaction with the electric field vector E in the radiation of the beam 24. Radiation having E and H components perpendicular to the axis 36 propagates in the direction of arrow 40 parallel to the axis 36. With reference also to Figures 12-16, the interaction between the H component and the filter elements 38 is dependent on the angle of incidence between the rays of radiation and normal to the lamina surface. Figure 13 shows a nonzero angle of incidence for a wave of radiation propagating in a direction, indicated by the arrow 40, which is inclined relative to the normal to the front surface 32, the inclination being in a plane containing the direction of the magnetic field vector H. The interaction is negligibly small for a zero angle of incidence, and increases with increasing angle of incidence. The interaction with the H component is characterized by an inducing of an electric current within each filter element 38 and a consequential dissipation of energy within each filter element 38. The interaction therefore reduces the intensity of radiation propagating through the filter 30. The effect of the interaction with the H component is depicted in Figure 11 wherein the sidelobes 28 of the radiation pattern are shown by dashed lines while the main lobe 26 is shown by a solid line. The dashed lines indicate that the sidelobes 28 have been reduced in intensity by virtue of the foregoing interaction of the H component with the filter elements 38. It is noted that the sidelobes are directed in angles off boresight, in which case the radiation associated with each of the sidelobes 28 is incident at a nonzero incidence angle so that the foregoing interaction takes place for each of the sidelobes 28. However, with respect to the main lobe 26, there is essentially no interaction between the H component and the filter elements 38 because the filter 30 is essentially transparent to radiation propagating along the axis 36. Thereby, the filter 30 has provided significant improvement to the directive radiation pattern emanating from the dish 22 by a foregoing reduction in the strength of the sidelobes 28. While the foregoing improvement in radiation pattern has been demonstrated in the use of a radar antenna, it is to be understood that the angular filter 30 may also be used with other sources of radiation including antennas employed in microwave relay communication links. The arrangement of the array of filter elements 38 may be the same or different on successive ones of the laminae 34. In Figure 12, the array is presumed to be the same on each of the laminae 34 with an element 38 on the lamina 34 at the back of the filter 30 being in line with the corresponding element 38 on the lamina 34 at the front of the filter 30. In Figure 12, pieces of the front and middle laminae 34 have been cut away to show the placement of the elements 38 on the front surfaces of each of the laminae 34. The spacing between the surfaces of the laminae 34 is indicated by the letter z; the spacing on centers between the elements 38 in the horizontal and vertical directions are indicated, respectively, by the letters x and y. Each of the elements 38 may be formed in accordance with the technology of printed-circuit construction wherein each of the elements 38 is formed as a deposit of an electrically conducting material such as copper. The width, w, and depth, d, can be chosen to provide the desired amount of resistance around the loop of the element 38. The amount of resistivity can also be selected by use of other materials such as carbon. Alternatively, the resistance can be provided by a specific resistor inserted in series with a loop of high conductivity. Thus, the resistance may either be continuous along the loop or lumped at one or more points within the loop. The spacing of the elements 38, as indicated by the dimensions x and y is preferably less than one-half wavelength so that the elements 38 appear as a continuum of interactive elements to a wave of the radiation, rather than as individually dispersed sites of interaction. It is also noted that the inductance of a loop of the element 38 is also dependent on the diameter, a, width, w, and depth, d, dimensions shown in Figures 13, 14, 15. Alternatively, each of the elements 38 may be configured as squares having sides of length, a, as shown in the elements 38A of Figure 15 instead of the elements 38 of Figure 14. Also, if desired, the sizes of the elements 38 may be decreased as shown by the smaller sized circular elements 38B of Figure 16 wherein the spacing of the elements has remained at approximately one-half wavelength. With the configuration of Figure 16, there is less interaction between the filter elements and the electric field component of the radiation. Also, the enclosed area of each of the elements 38B is smaller than the correspondng area of an element 38 resulting in reduced interaction with the magnetic field component of the radiation. Thus, the embodiment of Figure 16 has the advantage of reduced interaction with electric field at a cost of lesser attenuation of off axis radiation. With reference to Figures 17 and 18, an alternative embodiment of a filter element, designated 38C, provides for the introduction of capacitance in series with the flow of induced current around the loop of the element. The elements 38C comprises four members 42 of semcircular shape wherein two members 42 are disposed on one side of a lamina 34, and the other two members 42 are disposed on the opposite side of the lamina 34 in registration with the first set of two members 42. In each set of the two members 42, the members 42 are spaced apart by gaps 44. The two sets of members 42 are disposed with the respective gaps 44 of each set being staggered so that the gap 44 of one step lies opposite a member 42 of the other set. With this arrangement the two sets of members with a thin layer 34A (Figure 18) of the material of the lamina 34 therebetween constitute the filter element 38C. If desired, the layer of material 34A may compose a dielectric other than that used in the fabrication of the lamina 34. The construction of the element 38C employs the well-known principles of stripline construction in which a succession of layers of material, both conducting and non-conducting, are built up on a substrate. Both the gaps 44 and the thickness of the layer 34A provide the necessary spacing between the members 42 to permit them to serve as the plates of a capacitor to current circulating in the loop. The capacitance in series with the inductance of the loop provides a resonant enhancement of the circulating loop current without enhancing the unwanted interaction with the electric field of the wave. This increases the attenuation of off-axis radiation without increasing attenuation at normal incidence. With reference to Figures 19-23, there is a showing of further embodiments of filter elements which provide for the inclusion of one or more of the characteristics of resistance, capacitance, and electric-field shielding. Figure 19 corresponds to a loop of the element 38 wherein the loop is fabricated of electrically conducting material having little or no resistance, and a resistor 46 is inserted in series with the loop at a specified point. Also provided is an electric-field shield composed of arcuate electrically-conductive strips 48 which are located at ± 90 o from the resistor location, are electrically insulated from the loop 51 of the filter element, and are electrically connected together by a conductor 52 formed as a strip embedded within material of a lamina 34 and spaced apart from the loop 51 so as to be insulated therefrom. This combination of resistor and shield reduces the harmful interaction with electric field. In Figure 20, there is shown an alternative form of shielding accomplished by means of an electrical conductor 54 formed as a strip within the plane of the loop 51 and connected thereto between a pair of diametrically opposed points. Resistors 46 are disposed in each half of the conducting loop 51 midway between the strip connection points on the loop. This combination of conductor and resistors also reduces the harmful interaction with electric field. In Figure 21, the conducting loop 51 is shown having resistor 46 in series as well as capacitor 56 in series, which capacitor can be provided by the gap structure disclosed in Figures 17 and 18. With the structure of Figure 21, a resonance is introduced between the capacitor 56, and the inherent inductance in the conductor of the loop 51. This resonance tends to accentuate the interaction of the magnetic field component H without introducing any additional interaction with the electric field component E. If desired, the filter elements can be constructed of smaller size with the arrangement of Figure 21, thereby reducing the interaction with the electric field while maintaining the desired magnetic-field interaction and power dissipation by virtue of the resonance effect. In Figure 22, the structure of Figure 21 has been combined with an electric field shield such as that disclosed in Figure 19, which shield comprises the strips 48 and the interconnecting conductor 52. Thereby, the beneficial features of the filter associated with both the shielding effect and the resonance effect, respectively of Figures 19 and 21, have been combined in the single structure of Figure 22. The combination of shielding and resonance is also shown in the structure of Figure 23 wherein the shielding of Figure 20, composed of the conductor 54, is combined with the resonance associated with the capacitors 56 and the symmetrical construction of Figure 10. Thus, Figure 23 shows in each branch of the loop 51, by way of example, a resistor 46 and two capacitors 56, the capacitors 56 being associated with the structure disclosed in Figures 17 and 18 to provide a resonance between the inherent inductance of the conductor of the loop 51 in cooperation with the capacitance associated with the gaps and the spacing between the opposed sets of the members 42 of Figures 17-18. In Figure 3, the preferred curve shows the effect of the interaction of the magnetic field component with filter elements 38. As has been noted above, the interaction results in the inducing of a current within the loop 51 with an associated dissipation of power produced by the passage of current through a resistance. Such power dissipation is proportional to the square of the value of current, with the value of current itself being dependent on approximately the sine of the angle of incidence. The attenuation resulting from the dissipation of power from an off-boresight electromagnetic wave is portrayed in the graph of Figure 3 wherein the vertical axis, plotted in decibels, has been normalized with respect to the frequency of the radiation. The normalization is obtained by dividing the value in decibels by the wavelength as indicated adjacent the vertical axis of the graph. The horizontal axis is scaled in degrees of angle of incidence. The resulting attenuation, shown as the preferred trace is small at normal incidence (0 o ) and is characterized by a relatively slow change at low angles of incidence, a more rapid change in median ranges of angle of incidence, and then a relatively slow change at still larger angles of incidence. The relatively slow change at low angles of incidence is useful in the case of directive antennas wherein the beamwidth is several degrees or less, and wherein a troublesome sidelobe is, possibly, as much as 30 o off of boresight. As shown in the graph of Figure 3, such a sidelobe would be substantially attenuated while the main lobe would remain substantially unchanged by the filter 30. In the construction of the invention of Figures 11-23, the filter may be untuned, or it may be tuned to a desired frequency band for enhanced attenuation by addition of capacitance to the filter elements 38. In addition, the amount of resistance in a loop 50 of a filter element 38 can be selected for a maximum amount of power dissipation by the loop current. In addition, the filter 30 may be viewed as a medium which attenuates an electromagnetic signal propagating therethrough. The foregoing parameters, accordingly, are useful in the design of the filter of the invention or operation in a specific environment, such as with the radar antenna 20 of Figure 11. The foregoing description has provided for the construction of an angular filter, in accordance with the invention, wherein off-boresight propagation of electromagnetic waves is attenuated in favor of an electromagnetic wave propagating along the boresight axis by the mechanism of interaction of the magnetic field component of the electromagnetic waves with the loop-type elements of the angular filter. In addition, the foregoing construction has minimized reflection of the electric field component of the electromagnetic wave from the elements of the filter.

Claims: A filter (50,30) for filtering an electromagnetic wave energy signal incident thereon, said filter comprising: a substantially planar array of resistive elements (100,38) forming a substantially non-reflective array and being capable of interacting with an incident electromagnetic wave; and support means (200,34) formed of dielectric material, said support means supporting said elements in said substantially planar array and being substantially transparent to an incident electromagnetic wave; characterized in that: said resistive elements (100,38) are adapted such that an electromagnetic wave incident on said filter in a direction (300) normal to said planar array passes through said filter and said resistive elements (100,38) are adapted to each dissipatively attenuate an electromagnetic wave incident on said filter at an angle to said normal direction (300). A filter according to claim 1 characterized in that said resistive elements (100) are each elongate along an axis and are supported parallel to one another with their axes normal to said planar array, said filter (50) having an axial loss tangent for a given frequency of electromagnetic energy in the range of from 0.5 to 2.0, said axial loss tangent being defined as the axial conductance of the filter divided by the given frequency in radians per second and divided by the permittivity of free space. A filter according to claim 2 characterized in that said axial loss tangent is substantially equal to unity. A filter according to any one of claims 1 to 3 characterized in that said array has a square lattice. A filter according to any one of claims 1 to 4 characterized in that said support means comprises sheets of dielectric material (200). A filter according to claim 5 characterized in that said resistive elements (100) are screen printed on said sheets of dielectric material (200), said sheets being stacked and extending in planes normal to said planar array. A filter according to claim 6 characterized in that said sheets of dielectric material (200) have spaces therebetween. A filter according to claim 6 characterized in that said sheets of dielectric material (201) have spacers (202) of dielectric foam material therebetween. A filter according to claim 1 characterized in that said resistive elements (38) are loops disposed parallel to one another substantially in a plane in said substantially planar array; each said element (38) being surrounded by said dielectric material support means (34) and being held in a preset position in said array by said support means; each said element (38) being electrically independent of the other elements; each said element (38) comprising an electrically conductive member (51) curved in said plane for interaction with the magnetic vector component of a wave incident on said filter at an angle to said normal direction, there being substantially no interaction between said elements (38) and said magnetic vector for a wave incident on the filter in the normal direction, whereby the amount of said interaction and a consequent attenuation of wave energy increases with an increasing angle of incidence away from the normal direction. A filter according to claim 9 characterized in that each said curved member (51) has the shape of an arc of a circle. A filter according to claim 10 characterized in that each said curved member (51) is circular. A filter according to claim 11 characterized in that said elements (38) are spaced apart by a centre to centre spacing greater than the diameter of said curved members (51). A filter according to claim 12 characterized in that said diameter is less than one quarter wavelength of an incident wave to reduce interaction between the electric vector component of said wave and said elements (38). A filter according to any one of claims 9 to 13 characterized in that each said element (38) comprises a plurality of said curved members (51) arranged along a closed path and spaced apart to form a capacitor for current induced in that element by an incident wave. A filter according to any one of claims 9 to 14 characterized in that each said element (38) further comprises a shielding element (48) for reducing interaction with the electric vector component of said wave. A filter according to claim 14 characterized in that said support means comprises laminae (34) of dielectric material, said members (51) being arranged in two groups spaced apart along said normal direction by one of said laminae. A filter according to claim 9 characterized in that said curved members (38A) are angled and are arranged in a rectangular pattern. A filter according to any one of claims 9 to 15 characterized by at least two said arrays disposed in substantially parallel said planes. A filter according to claim 18 characterized in that said arrays are flat planar arrays.

Inventors: HANNAN, PETER W.
Applicants: HAZELTINE CORPORATION
Application year: 1985
Main classification: H01Q15
IPC codes: H01Q15