CIRCUIT FOR DETECTING LEVEL OF INPUT VOLTAGE

Abstract: A circuit for detecting the level of an input voltage is disclosed, in which the input voltage is converted into a current and this current is compared with a reference current obtained by converting a reference voltage. To accomplish such circuit functions, this level detection circuit comprises means for converting the input voltage into a first current, means for converting the reference voltage into a second current, means for producing a third current relative to the first current, means for producing a fourth current relative to the second current, and means for combining the third and fourth current. A detection output terminal is coupled to the combining means.

Description: The present invention relates to a circuit for detecting a level of an input voltage, and particularly to a level detection circuit composed of insulated gate field effect transistors. A level detection circuit employing a ratio circuit and an inverter circuit has been proposed to detect the level of an input voltage, particularly to detect that the input voltage is larger in absolute value than a power supply voltage. The ratio circuit includes first and second insulated gate field effect transistors (IGFET's) connected in series between power supply terminals. The first IGFET is of an enhancement type and supplied with the input voltage, and the second IGFET is of a depletion type. The connection point of the first and second IGFET's is an output terminal of the ratio circuit to which the inverter circuit is connected. The output level of the ratio circuit is changed in accordance with the ratio of the internal impedances of the first and second IGFET's and the internal impedance of the first IGFET is controlled by the input voltage. When the output level of the ratio circuit exceeds the threshold level of the inverter circuit, the inverter circuit changes its output logic level. Thus, the detection level of the input voltage is determined by the internal impedances of the first and second IGFET's and the threshold level of the inverter circuit. Accordingly, various detection levels of the input voltage can be designed with facility, and thus it is possible to generate the detection output when the input voltage exceeds the power supply voltage. However, precise control of the threshold level of the inverter circuit is very difficult. Moreover, the internal impedances of the IGFET's depend on the threshold levels thereof, and it is also quite difficult to precisely determine the threshold levels of the IGFET's. As a result, the detection level of the input voltage deviates considerably from a designed value. Therefore, an object of the present invention is to provide an improved detection circuit for detecting the level of an input voltage. Another object of the present invention is to provide a level detection circuit in which affection of the detection level of an input voltage upon deviation or variation of threshold levels of IGFET's is minimized. A level detection circuit according to the present invention comprises an input terminal supplied with an input voltage, a reference terminal supplied with a reference voltage, first means coupled to the input terminal for converting the input voltage into a first current, second means coupled to the reference terminal for converting the reference voltage into a second current, third means coupled to the first means for producing a third current relative to the first current, fourth means coupled to the second means for producing a fourth current relative to the second current, fifth means coupled to the third and fourth means for combining the third and fourth currents, and an output terminal coupled to the fifth means. In the case where the input voltage is smaller in absolute value than the reference voltage, the first current is smaller than the second current. Assuming that the ratio of the third current to the first current and that of the fourth current to the second current are 1, respectively, the third current is also smaller than the fourth current. The output terminal thereby takes a first logic level. When the input voltage becomes larger in absolute value than the reference voltage, the first current becomes larger than the second current. The third current thereby takes a value larger than the fourth current. As a result, the output terminal is inverted from the first logic level to a second logic level. Thus, the circuit according to the present invention detects the level of the input voltage by comparing the currents relative to the input voltage and the reference voltage. Such a current comparison circuit is stabilized against the deviation of and the change in threshold levels of IGFET's and the change in power supply voltage. The detection level of the input voltage is controlled by the current convertion ratio between the first and second means and/or the current producing ratio between the third and fourth means. Accordingly, the various detection levels can be designed facilely without deviation from the designed value. It is convenient to constitute the third and fourth means with a current mirror circuit. The current mirror circuit is composed of two IGFET's and produces an output current proportional to an input current, and the ratio of the output current to the input current is determined by the ratio in conductances between those IGFET's. Each of the first and second means may be composed of an IGFET having a gate supplied with the input voltage or the reference voltage. The invention will now be described, by way of example, with reference to the accompanying drawings, in which Fig. 1 is a circuit diagram showing a level detection circuit according to prior art; Fig. 2 is a graph representing a waveform of an output voltage V OUT with respect to an input voltage V IN in the circuit shown in Fig. 1; Fig. 3 is a circuit diagram representing an embodiment of the present invention; Fig. 4 is a graph for explaining an operation of the circuit shown in Fig. 3; Fig. 5 is a graph representing a waveform of an output voltage V OUT with respect to an input voltage V IN in the circuit shown in Fig. 3; and Fig. 6 is a circuit diagram showing another embodiment of the present invention. A level detection circuit according to prior art is shown in Fig. 1 to facilitate the understanding of the present invention. This circuit is composed of an enhancement MOS (Metal-Oxide-Semiconductor) transistor Q₁ of an N-channel type, a depletion MOS transistor Q₂ of the N-channel type, and an inverter 2. The transistors Q₁ and Q₂ are connected in series between power supply terminals V cc and GND. The gate of the transistor Q₁ is connected to an input terminal 1 supplied with an input voltage V IN to be detected, and the gate of the transistor Q₂ is grounded. The inverter 2 is composed of a P-channel MOS transistor Q₃ and an N-channel MOS transistor Q₄ connected in series between the power supply terminals V cc and GND. The gates of the transistors Q₃ and Q₄ are connected in common to the node N₁, and an output terminal 3 is connected to the connection point of the transistors Q₃ and Q₄. In the case where the input voltage V IN is in a low level, i.e., in a ground level, the transistor Q₁ is in a nonconductive state. On the other hand, the transistor Q₂ is of the depletion type, and hence the node N₁ takes the ground level. As a result, the transistor Q₃ is turned ON and the output voltage V OUT takes a high level, i.e., a power supply voltage V cc . When the input voltage V IN rises to the threshold level of the transistor Q₁, the transistor Q₁ is brought into a conductive state. The potential at the node N₁ is thus determined by the ratio of the internal impedance of the transistor Q₁ to that of the transistor Q₂. The internal impedance of the transistor Q₁ is considerably high at the time when the input voltage V IN rises to the threshold level of the transistor Q₁, and therefore the potential at the node N₁ is near substantially to the ground level. The output voltage V OUT at the terminal 3 is thereby maintained to the high level. The internal impedance of the transistor Q₁ is lowered by the increase in level of the input voltage V IN , and the potential at the node N₁ is increased in response thereto. Accordingly, when the input voltage V IN increases to such a level that the potential at the node N₁ exceeds the threshold level of the inverter 2, the transistor Q₄ is turned ON to invert the output voltage V OUT to the low level. Thus, the detection level of the input voltage V IN at which the output voltage V OUT changes from the high level to the low level, is determined by the internal impedance of the transistors Q₁ and Q₂ and the threshold level of the inverter 2. Accordingly, various detection levels of the input voltage V IN can be designed by adjusting these parameters. In some of IC's (integrate circuit devices), there is further provided a test circuit for testing an operation of an internal signal processing circuit. In this case, the test circuit should not be activated in a normal operation state of the IC, and further one of the external lead terminals supplied with signals to be processed by the internal signal processing circuit is used in common as a terminal supplied with a testing voltage for activating the test circuit. The signal to be processed has a level amplitude changing between the power voltage V CC and the ground voltage GND. The testing voltage is therefore selected to take a level larger than the power voltage V CC , whereby the testing voltage is distinguished from the signal to be processed. Accordingly, such a circuit is required, that produces an output signal in response only to the application of the testing voltage. The circuit shown in Fig. 1 can be utilized for detecting the testing voltage. More specifically, the internal impedances of the transistors Q₁ and Q₂ are selected such that the potential V N1 at the node N₁ exceeds the threshold level V TH2 of the inverter 2 when the input voltage V IN is equal to or larger than 7 V, as shown in Fig. 2. The power supply voltage V CC is 5 V. In the normal operation, the voltage at the input terminal 1 does not become larger than the power supply voltage V CC , and therefore the output voltage V OUT is maintained to the high level, as apparent from Fig. 2. On the other hand, the testing voltage for activating the internal test circuit has a level twice as large as the power supply voltage V cc , 10 V. Accordingly, in response to the application of the testing voltage to the terminal 1, the potential V N1 at the node N₁ exceeds the threshold level V TH2 of the inverter 2 to invert the output voltage V OUT from the high level to the low level. The output terminal 3 is coupled to the internal test circuit, so that the test circuit is activated to test the operation of the internal signal processing circuit. However, it is very difficult to precisely control the internal impedances of the transistors Q₁ and Q₂. Particularly, the transistor Q₁ is of an enhancement type and the transistor Q₂ is a depletion type, and therefore the deviation from the designed impedance of the transistor Q₁ occurs independently of the deviation from the designed impedance of the transistor Q₂. As a result, the rate of change in the potential V NI at the node N₁ is unavoidably deviated from the designed rate V N1-1 , which is represented by V N1-2 or V N1-3 in Fig. 2. The level of the input voltage V IN at which the output voltage V OUT changes to the low level is thereby shifted to the level V IN-2 or V IN-3 , as shown in Fig. 2. If the level V IN-2 is larger than 10 V, the testing voltage is not detected. In the case where the level V IN -3 is smaller than the power voltage V CC , the output voltage V OUT taking the low level is produced even when the IC attains the normal operation. The threshold level V TH2 of the inverter 2 is also deviated from the designed value, as represented by V TH2-1 or V TH2-2 in Fig. 2, resulting in change in level of the input voltage V IN at which the output voltage V OUT is inverted to the low level. As described above, the detect level of the circuit according to prior art is not stabilized against the deviations from the designed impedances of the transistors Q₁ and Q₂ and from the designed threshold level of the inverter 2. Referring to Fig. 3, there is shown a level detection circuit according to an embodiment of the present invention. This circuit is composed of complementary MOS transistors. An input voltage V IN to be detected is supplied to a first input terminal 10 which is in turn connected to the gate of an N-channel MOS transistor Q₁₂. The transistor Q₁₂ is connected between a ground terminal and a first node N₁₁, and thus converts the input voltage V IN into a first current I₁ relative to the level of the input voltage V IN . A second input terminal 11 receives a reference voltage V REF and is connected to the gate of an N-channel MOS transistor Q₁₆ connected between the ground terminal and a third node N₁₃. The reference voltage V REF is converted into a second current I₂ by the transistor Q₁₆. The first current I₁ is supplied as an input current to a first current mirror circuit 15 composed of two P-channel MOS transistors Q₁₁ and Q₁₃. The transistor Q₁₁ is connected between the node N₁₁ and a power voltage supply terminal V CC , and the gate thereof is connected to the node N₁₁ and further connected to the gate of the transistor Q₁₃. The transistor Q₁₃ is connected between the power terminal V CC and a second node N₁₂. Thus, the gates and sources of the transistors Q₁₁ and Q₁₃ are connected in common, respectively, and the gate of the transistor Q₁₁ is connected to its drain. Therefore, the first current mirror circuit 15 produces a third current I₃ proportional to the first current I₁. Assuming that the conductances of the transistors Q₁₁ and Q₁₃ are G m11 and G m13 , respectively, the third current I₃ is as follows: I₃ = I₁ × (G m13 /G m11 ) (1) The second current I₂ is supplied to a second current mirror circuit 16 composed of two P-channel MOS transistors Q₁₅ and Q₁₇. The transistor Q₁₅ is connected between the power terminal V CC and the third node N₁₃, and the gate thereof is connected to the node N₁₃. The gate of the transistor Q₁₅ is further connected to the gate of the transistor Q₁₇ connected between the power terminal V CC and a fourth node N₁₄. The second current mirror circuit 16 thus produces a fourth current I₄ proportional to the second current I₂. When the conductances of the transistors Q₁₅ and Q₁₇ are G m15 and G m17 , respectively, the fourth current I₄ takes the following value: I₄ = I₂ × (G m17 /G m15 ) (2) The fourth current I₄ is supplied to a third current mirror circuit 17 as its input current, which is composed ot two N-channel MOS transistors Q₁₈ and Q₁₄. The transistor Q₁₈ is connected between the fourth node N₁₄ and the ground terminal. The gate of the transistor Q₁₈ is connected to the node N₁₄ and further connected to the gate of the transistor Q₁₄ which is in turn connected between the second node N₁₂ and the ground terminal. Assuming that the conductances of the transistors Q₁₄ and Q₁₈ are G m14 and G m18 , respectively, the third current mirror circuit 17 responds to the fourth current I₄ and produces a fifth current I₅ taking the following value: I₅ = I₄ × (G m14 /G m18 ) (3( The second node N₁₂ is further connected to an output terminal 12 from which a detection output V OUT is derived. All the transistors Q₁₁ to Q₁₈ are of an enhancement type. In operation, the transistor Q₁₆ responds to the reference voltage V REF and produces the second current I₂ relative to the level of V REF . Assuming that G m15 = G m17 and that G m18 = G m14 , the fourth and fifth currents I₄ and I₅ become equal to the second current I₂, as apparent from the equations (2) and (3). In other words, the same current as the second current I₂ flows into the transistors Q₁₅, Q₁₇ and Q₁₈ respectively, and the transistor Q₁₄ is biased so as to draw the current equal to the second current I₂ from the node N₁₂. As a result, the characteristic of the drain current I DN to the drain-source voltage V DS of the transistor Q₁₄ is represented by a line 100 in Fig. 4. On the other hand, the input voltage V IN is converted into the first current I₁ by the transistor Q₁₂. Assuming that the conductances of the transistors Q₁₂ and Q₁₆ are equal to each other, the first current I₁ becomes smaller than the second current I₂ when the input voltage V IN is smaller than the reference voltage V REF . If the conductance of the transistor Q₁₁ is equal to that of the transistor Q₁₃ (G m11 = G m13 ), the third current I₃ becomes equal to the first current I₁. To the contrary, the transistor Q₁₄ has the ability to draw the current equal to the second current I₂. Accordingly, the characteristic of the drain current I DP to the drain-source voltage V DS of the transistor Q₁₃ is shown by a line 110 in Fig. 4. As apparent from Fig. 4, the line 110 intersects with the line 100 in the unsaturated region of the transistor Q₁₄. As a result, the voltage level V OUT1 at the intersection of the lines 100 and 110 is produced as the output voltage V OUT . The level of V OUT1 is substantially the low level, i.e., the ground level as shown in Fig. 4. In other words, since the transistor Q₁₃ has only the ability to supply the current smaller than the second current I₂ which the transistor Q₁₄ can draw, the drain-source voltage of the transistor Q₁₄ is lowered, so that the output voltage V OUT takes the V OUT1 level. In accordance with the increase in level of the input voltage V IN , the current I₁ (i.e., I₃) is increased, so that the intersection of the lines 100 and 110 changes along the line 100. However, so long as the input voltage V IN is smaller than the reference voltage V REF , the intersection of the lines 100 and 110 exists in the unsaturated region of the transistor Q₁₄. Accordingly, the output voltage V OUT changes only from 0 V to 0.5 V. These levels are regarded as the low level. When the input voltage V IN becomes equal to the reference voltage V REF , the currents flowing through all the transistors Q₁₁ to Q₁₈ are equal to one another. Accordingly, the lines 100 and 110 overlap with each other in their straight portions. As a result, the output voltage V OUT takes an intermediate level between the power voltage V CC and the ground voltage GND. When the input voltage V IN becomes larger even slightly than the reference voltage V REF , the first current I₁ takes a value larger than the second current I₂. The current supply ability of the transistor Q₁₃ is thereby made larger than the current draw ability of the transistor Q₁₄. Accordingly, the characteristic of the drain current I DP to the drain-source voltage V DS of the transistor Q₁₃ is changed from the line 110 to a line 120 as shown in Fig. 3. In other words, the intersection of the lines 100 and 120 exists in the unsaturated region of the transistor Q₁₃. As a result, the output voltage V OUT takes a second level V OUT2 which is near substantially to the power supply voltage V CC . The difference between the second level V OUT2 and the power voltage V CC is only 0.5 V in a maximum value as shown in Fig. 4, and it is decreased by the increase of the input voltage V IN . Therefore, the second level V OUT is regarded as the high level. As described above, assuming that G m11 = G m13 , G m12 = G m16 , G m14 = G m18 , and G m15 = G m17 , the output voltage V OUT changes the low level to the high level, when the input voltage V IN exceeds the reference voltage V REF . The ratio in conductances of two transistors in pair, Q₁₁ and Q₁₃, Q₁₂ and Q₁₆, Q₁₄ and Q₁₈, and Q₁₅ and Q₁₇, is controlled by the geometrical dimensions of the respective transistors. The respective conductances of the transistors Q₁₁ to Q₁₈ may be changed by the deviation from the designed threshold levels of the transistors Q₁₁ to Q₁₈. However, two transistors in pair, Q₁₁ and Q₁₃, Q₁₂ and Q₁₆, Q₁₄ and Q₁₈, and Q₁₅ and Q₁₇, are of the same channel type, and both of the transistors in pair are of an enhancement type. Accordingly, if the threshold level of one of the transistors in pair is deviated from the designed value, the threshold level of the other transistor is also deviated from the designed value by the same value. The ratio in conductances of two transistors in pair is thereby maintained constantly. As a result, if the line 100 is shifted to a line 101 or 102, the line 110 is shifted to a line 111 or 112, respectively, by the same value. The line 120 is also shifted to a line 121 or 122, respectively. In other words, the ratio of the current convertion rate of the input voltage V IN by the transistor Q₁₂ to that of the reference voltage V REF by the transistor Q₁₆ is stabilized, and the ratios between the currents I₁ and I₃, between those I₂ and I₄, and between those I₄ and I₅ are also stabilized. Moreover, the detection level of the input voltage V IN can be changed arbitrary by adjusting the ratio in conductances of the transistors in pair. For example, by selecting the ratio in conductances such that G m12 < G m16 , G m11 > G m13 G m14 > G m18 , or G m17 > G m15 , the detection level of the input voltage V IN becomes larger than the reference voltage V REF . The circuit shown in Fig. 3 is also utilized for detecting the afore mentioned testing voltage. In this case, the second input terminal 11 is connected to the power supply terminal V CC , and the ratio of the conductance of the transistor Q₁₄ to that of the transistor Q₁₈ is selected to 1.5 : 1. The ratios in conductances between the transistors Q₁₁ and Q₁₃, Q₁₂ and Q₁₆, and Q₁₅ and Q₁₇ are designed to be 1, respectively. According to these circuit designs, even when the input voltage V IN takes the power supply voltage V CC , the current draw ability of the transistor Q₁₄ is larger than the current supply ability of the transistor Q₁₃, so that the output voltage is maintained to the low level, as shown in Fig. 5. When the input voltage V IN increases to 7 V, the output voltage V OUT starts to change its level. The output voltage V OUT is inverted quickly from the low level to the high level, when the input voltage V IN exceed 7 V. Accordingly, the testing voltage is detected. The level of the input voltage V IN at which the output voltage is inverted, 7 V, is stabilized against the deviation from the designed threshold level of each transistor, as already mentioned. The input voltage V IN and the reference voltage V REF can be supplied to the second terminal 11 and the first terminal 10, respectively. In this case, the output voltage V OUT is changed from the high level to the low level, when the input voltage V IN exceeds a predetermined detection level. Fig. 4 shows another embodiment of the present invention, in which the same constituents as those shown in Fig. 2 are denoted by the same references. This circuit further includes two P-channel MOS transistors Q₁₉ and Q₂₂, and four N-channel MOS transistors Q₂₀, Q₂₁, Q₂₃ and Q₂₄, and each of these transistors is of an enhancement type. The transistor Q₁₉ is connected between the power terminal V CC and the first node N₁₁, and the gate thereof is connected to a control terminal 13. The transistor Q₂₂ is connected between the power supply terminal V CC and the fourth node N₁₄, and the gate thereof is connected to the control terminal 13. The transistor Q₂₀ has its gate connected to the control terminal 13 and is inserted between the transistor Q₁₂ and the ground terminal. The transistor Q₂₁ is inserted between the transistor Q₁₄ and the ground terminal. The gate of the transistor Q₂₁ is supplied with the power voltage V CC . The transistor Q₂₃ is inserted between the transistor Q₁₆ and the ground terminal and has its gate connected to the control terminal 13. The transistor Q₂₄ is connected between the transistor Q₁₈ and the ground terminal, and the gate of the transistor Q₂₄ is connected to the control terminal 13. The control terminal 13 is supplied with a control signal ø. The circuit shown in Fig. 4 is switched over between the stand-by mode and the activated mode by the control signal ø. More specifically, when this circuit is brought into the stand-by mode, i.e., when the control signal ø takes the low level, the transistors Q₁₉ and Q₂₂ are turned ON, whereas the transistors Q₂₀, Q₂₃ and Q₂₄ are turned OFF. The transistors Q₁₁ and Q₁₃ are thereby brought into the nonconductive state. The transistor Q₁₄ is turned ON by the transistor Q₂₂, and the transistor Q₂₁ is in the conductive state. Accordingly, this circuit does not carry out the level detection operation, and the output voltage V OUT is held to the low level. Moreover, the transistors Q₂₀, Q₁₃, Q₂₃ and Q₂₄ are in the nonconductive state, and therefore no d.c. current path is formed between the power supply terminals V CC and GND, so that a power dissipation in the stand-by mode is remarkably reduced. When the control signal ø changes to the high level, the transistors Q₁₉ and Q₂₂ are turned OFF, and the transistors Q₂₀, Q₂₃ and Q₂₄ are turned ON. As a result, this circuit is in the activated state to carry out the level detection operation, similarly to the circuit shown in Fig. 2. It is preferable that the conductances of the transistors Q₂₀, Q₂₁, Q₂₃ and Q₂₄ are sufficiently high to neglect the insertion of these transistors. The transistor Q₂₁ may be omitted, because it is maintained in the conductive state during the stand-by mode as well as the activated mode. However, in order to obtain the symmetry between a current path including the transistors Q₁₈ and Q₂₄ and that including the transistor Q₁₄, the transistor Q₂₁ is provided. As already mentioned, the second input terminal may be connected to the power terminal V CC . In this case, the detection output is produced when the input voltage V IN exceeds the power voltage V CC , and therefor this circuit can be applied to a test circuit. This circuit is also applicable to a write circuit in an electrically programmable read only memory device (EPROM). The input voltage V IN and the reference V REF may be supplied to the second terminal 11 and the first terminal 10, respectively. The present invention is not limited to the above- mentioned embodiments, but may be modified and changed without departing the scope and spirit of the invention. For example, the channel types of the respective transistors may be changed.

Claims: 1. A level detection circuit comprising an input terminal supplied with an input voltage, a reference terminal supplied with a reference voltage, first means coupled to said input terminal for converting said input voltage into a first current, second means coupled to said reference terminal for converting said reference voltage into a second current, third means coupled to said first means for producing a third current relative to said first current, fourth means coupled to said second means for producing a fourth current relative to said second current, fifth means coupled to said third and fourth means for combining said third and fourth currents, and an output terminal coupled to said fifth means. 2. The circuit as claimed in claim 1, wherein said fifth means controls the potential at said output terminal in response to said third and fourth currents. 3. The circuit as claimed in claim 1, wherein said fifth means comprises a current mirror circuit having an input end connected to said third means and an output end connected to said fourth means, said output terminal being coupled to the output end of said current mirror circuit. 4. The circuit as claimed in claim 1, wherein said fifth means comprises a current mirror circuit having an input end connected to said fourth means and an output end connected to said third means, said output terminal being coupled to the output end of said current mirror circuit. 5. A comparator comprising first and second input terminals, means for supply an input voltage to one of said first and second input terminals, means for supplying a reference voltage to the other of said first and second input terminals, means coupled to said first input terminal for converting a voltage supplied to said first input terminal into a first current, means coupled to said second input terminal for converting a voltage supplied to said second input terminal into a second current, a first current mirror circuit responding to said first current and producing a third current relative to said first current, a second current mirror circuit responding to said second current and producing a fourth current relative to said second current, a third current mirror circuit having an input end supplied with said third current and an output end supplied with said fourth current, and an output terminal coupled to the output end of said third current mirror circuit. 6. The comparator as claimed in claim 5, wherein a power supply voltage is used as said reference voltage. 7. The comparator as claimed in claim 5, wherein each of said first to third current mirror circuits includes first and second field effect transistors, the gates of said first and second transistors being connected in common, and the gate and drain of said first transistor being connected in common. 8. A transistor circuit comprising first and second power supply terminals, first and second input terminals, an output terminal, first to fourth node, a first transistor connected between said first power supply terminal and said first node, means for connecting said first input terminal to the control electrode of said first transistor, a first current mirror circuit having an input end coupled to said first node, an output end coupled to said second node and a common end coupled to said second power supply terminal, a second transistor connected between said first power supply terminal and said third node, means for connecting said second input terminal to the control electrode of said second transistor, a second current mirror circuit having an input end coupled to said third node, an output end coupled to said fourth node and a common end coupled to said second power supply terminal, a third current mirror circuit having an input end coupled to said fourth node, an output end coupled to said second node and a common end coupled to said first power supply terminal, means for connecting said output terminal to said second node, means for supplying an input voltage to one of said first and second input terminals, and means for supplying a reference voltage to the other of said first and second input terminals. 9. The circuit as claimed in claim 8, wherein said other of said first and second input terminal is connected to said second power supply terminal to use a potential at said second power supply terminal as said reference voltage. 10. The circuit as claimed in claim 8, wherein said first current mirror circuit includes a third transistor connected between said first node and said second power supply terminal and a fourth transistor connected between said second node and said second power supply terminal, the control electrode of said third transistor being connected to said first node and the control electrode of said fourth transistor, said second current mirror circuit including a fifth transistor connected between said third node and said second power supply terminal and a sixth transistor connected between said fourth node and said second power supply terminal, the control electrode of said fifth transistor being connected to said third node and to the control electrode of said sixth transistor, said third current mirror circuit including a seventh transistor connected between said fourth node and said first power supply terminal and a eighth transistor connected between said second node and said first power supply terminal, and the control electrode of said seventh transistor being connected to said fourth node and to the control electrode of said eighth transistor. 11. The circuit as claimed in claim 10, wherein said first, second, seventh and eighth transistors are of one conductivity type, and said third, fourth, fifth and sixth transistors are of opposite conductivity type. 12. A transistor circuit comprising first and second potential terminals, first and second input terminals, first to fourth nodes, a first field effect transistor connected between said first potential terminal and said first node and having a gate connected to said first terminal, a second field effect transistor connected between said first potential terminal and said second node and having a gate connected to said second input terminal, a third field effect transistor connected between said second potential terminal and said first node and having a gate connected to said first node, a fourth field effect transistor connected between said second potential terminal and said third node and having a gate connected to the gate of said third transistor, a fifth field effect transistor connected between said second potential terminal and said second node and having a gate connected to said second node, a sixth field effect transistor connected between said second potential terminal and said fourth node and having a gate connected to the gate of said fifth transistor, a seventh transistor connected between said fourth node and said first potential terminal and having a gate connected to said fourth node, a eighth transistor connected between said third node and said first potential terminal and having a gate connected to the gate of said seventh transistor, means for supplying an input voltage to one of said first and second input terminals, means for supplying a reference voltage to the other of said first and second input terminal, and an output terminal coupled to said third node. 13. The circuit as claimed in claim 12, wherein said first, second, seventh and eighth transistors are of one channel type, and said third, fourth, fifth and sixth transistors are of opposite channel type. 14. A transistor circuit comprising first and second potential terminals, first to fourth nodes, a first series connection circuit of first and second field effect transistors provided between said first potential terminal and said first node, a first input terminal connected to the gate of said first transistor, a second series connection circuit of third and fourth field effect transistors provided between said first potential terminal and said second node, a second input terminal connected to the gate of said third transistor, a fifth field effect transistor connected between said second potential terminal and said first node and having a gate connected to said first node, a sixth field effect transistor connected between said second potential terminal and said third node and having a gate connected to the gate of said fifth transistor, a seventh field effect transistor connected between said second potential terminal and said second node and having a gate connected to said second node, an eighth field effect transistor connected between said second potential terminal and said fourth node and having a gate connected to the gate of said seventh transistor, a third series connection circuit of ninth and tenth field effect transistors provided between said third node and said first potential terminal, means for connecting the gate of said ninth transistor to said first potential terminal, a fourth series connection circuit of eleventh and twelfth field effect transistors provided between said fourth node and said first potential terminal, means for connecting the gate of said eleventh transistor to said fourth node and to the gate of said tenth transistor, an output terminal connected to said third node, a control terminal connected in common to the gates of said second, fourth and twelfth transistors, means for supplying an input voltage to one of said first and second input terminals, means for supplying a reference voltage to the other of said first and second input terminals, and means for supplying a control signal to said control terminal. 15. The circuit as claimed in claim 14, further comprising a thirteenth field effect transistor connected between said first node and said second potential terminal and a fourteenth field effect connected between said fourth node and said second potential terminal, said control terminal being further connected to the gates of said thirteenth and fourteenth transistors.

Inventors: KOBATAKE, HIROYUKI
Applicants: NEC CORPORATION
Application year: 1985
Main classification: G01R19
IPC codes: H03F3, G01R19, H03K5