MULTIPLE OUTPUT SWITCHING POWER SUPPLY

Abstract: A compact multiple output switching power supply having a series connection of a D.C. input power supply (E), a switching device (Q) and a primary winding (n p ) of a transformer (T₁) which supplies currents to loads during an on period of the switching device, and supplies constant output voltages V i (i = 1,2, ..., m) from m (m ≧ 2) secondary windings n i (i = 1,2 ..., m) of the transformer by turning on and off the switching device, includes means (T₂) for storing a magnetic energy during the on period of the switching device and discharging the stored energy to the loads during on off period, connected in series with the primary winding of the transformer. The output voltage V i (j = 2, 3, ..., m) is represented by where k j is a voltage drop in the circuit.

Description: BACKGROUND OF THE INVENTION The present invention relates to a multiple output switching power supply, and more particularly to a current type multiple output switching power supply suitable to supply a power to an OA or FA equipment which requires a high precision of output voltage. A current type switching power supply is disclosed in JP-A-55-120369 which discloses that a recovery current of a flywheel diode inserted in a secondary circuit of a transformer does not flow into a switching device in a primary circuit of the transformer and flow of an excessive current in an input circuit is suppressed even if the transformer is saturated by a surge voltage from the input circuit. However, it does not recognize a problem encountered in the multiple output switching power supply. JP-A-55-144775 discloses a technique to operate a switching device to stabilize one of the outputs so that the multiple outputs are stabilized. In the disclosed technique, a choke coil inserted in a secondary circuit of a transformer has a plurality of magnetically coupled windings. In the disclosed technique, it is not possible to determine the output voltages by turn ratios of a plurality of secondary windings wound on a transformer but turn ratios of the windings wound on the choke coil should also be taken into consideration. As a result, the more the number of outputs is, the more difficult to design the power supply. Further, since one output is stabilized by the operation of the switching device, variations of other output voltages due to a variation of the one output voltage, that is, voltage variations due to so called cross-regulation increase. Causes of the cross-regulation are leakage inductances and winding resistances of the transformer. In the disclosed technique, the leakage inductances and the winding resistances which cause the cross-regulation during an on period of the switching device affect to the transformer as well as leakage inductances and resistances of the windings of the choke coil and make it difficult to precisely control all of the output voltages. In a technique disclosed in JP-A-56-19119, an inductor is connected to an A.C. input. As a result, a ripple of a current flowing through the inductor is large, a current capacity of a rectifying/filtering circuit in a secondary circuit of a transformer is large and a maximum current flowing through a leakage inductor and a resistor of the transformer is large. As a result, outputs are affected by voltage drops thereacross and precision of the output voltages is lowered. A multiple output switching power supply with the features included in the first part of claim 1 is known from JP-A-57-177277. In that circuit, the voltage supplied to the load of only one of the secondary circuits is stabilized by controlling the ON-OFF operation of the switching device included in series connection at the primary side of the transformer, whereas each of the remaining secondary circuits requires its own voltage regulator circuit, each including a separate switching transistor and a controller. SUMMARY OF THE INVENTION It is an object of the present invention to provide a multiple output switching power supply which is capable of supplying constant voltages at a desired mutual ratio at the outputs of a plurality of secondary circuits with an uncomplicated circuit design. The invention meets this problem by the circuit characterized in claim 1. BRIEF DESCRIPTION OF THE DRAWINGS Fig. 1 shows a general multiple output switching power supply. Fig. 2 is an equivalent circuits of the circuit shown in Fig.1 during an on period of a transistor Q. Fig. 3 shows a circuit configuration of a prior art multiple output switching power supply having a plurality of windings on a choke coil. Fig. 4 is an equivalent circuit of Fig. 3. Fig. 5 shows an embodiment of the present invention. Fig. 6 is an equivalent circuit during an off period of the transistor Q, for illustrating an operation of the energy discharge circuit of Fig 5. Figs. 7 to 9 show other embodiments of the present invention. DESCRIPTION OF THE PREFERRED EMBODIMENTS A multiple output switching power supply is explained with reference to Fig. 1. E denotes a D.C. input power supply. A current I c flows, during an on period of a transistor Q which is a switching device, from a positive terminal of E through a reactor T₂ which stores a magnetic energy, a primary winding n p of a transformer T₁ to a negative terminal of E. Thus, voltages are developed across the primary winding n p and secondary windings n₁ and n₂ of the transformer T₁ with dotted ends being positive, and currents are supplied to loads from n₁ and n₂ through diodes D₁ and D₂, respectively. A capacitor C₁ smoothens the voltage supplied to the load connected to an output terminal 1 and it is charged with a voltage V₁. A capacitor C₂ also functions in a similar manner to C₁ and it is charged with a voltage V₂ which is supplied to the load connected to an output terminal 2. Numeral 10 denotes means which discharges, during an off period of the transistor Q, the magnetic energy stored in the reactor T₂ during the on period of the transistor Q. The operation of the circuit of Fig.1 is explained with reference to Fig. 2 which shows an equivalent circuit during the on period of the transistor Q. L₂ denotes an inductance of the reactor T₂, Z₁ denotes an impedance including a leakage inductance, a winding resistance and a circuit resistance of the winding n₁, and Z₂ denotes an impednace including a leakage inductance, a winding resistance and a circuit resistance of the winding n₂. The current flowing through the inductance L₂ shunts to the output terminals 1 and 2 with currents I₁ and I₂, respectively. The currents I₁ and I₂ meet the following relationship. I₁·Z₁ + V D1 + V₁ = I₂·Z₂ + V D2 + V₂ (1) where V D1 and V D2 are voltage drops across the diodes D₁ and D₂, respectively. From the formula (1), the volage V₂ at the output terminal 2 is expressed by V₂ = I₁·Z₁ + V D1 - (I₂·Z₂ + V D2 ) + V₁ (2) From the formula (2), it is seen that the voltage V₂ is represented by the voltage V₁ at the output terminal 1 and the circuit impedance. Referring to Fig. 1, the voltages induced across the windings of the transformer T₁ during the on period of the transistor Q is considered. The voltage induced across the winding n₁ with the dotted end being positive is equal to V₁ + V D1 , as seen from Fig. 1. Because of the voltage drop I₁·Z₁ due to the winding resistance and the circuit resistance of the transformer, the voltage V p shown by the following formula must be developed across the primary winding n p of T₁ with the dotted end being positive. If V p is given, a voltage V n2 induced across the winding n₂ with the dotted end being positive is expressed by From the formula (4), the voltage V₂ at the output terminal 2 is expressed by The voltage drop I₁·Z₁ + V D1 at the output terminal 1 is given by k₁ and the voltage drop I₂·Z₂ + V D2 at the output terminal 2 is given by k₂. Thus, from the formula (5), the voltage V₂ is expressed by From the above, it is seen that the voltage V₂ at the output terminal 2 is defined by the turn ratio of the secondary windings n₁ and n₂ of the transformer T₁, and the voltage V₁ at the output terminal 1 and the voltage drops in the circuit. Since k₁ and k₂ are generally smaller than V₁, V₂ can be expressed as follows without a large error. Thus, the turn ratios of the transformer T₁ can be substantially determined by the ratios of the output voltages and the circuit design is facilitated. In the above Figures, a multiple output switching power supply having two outputs was shown. When m outputs are included, the operation is similar and the output voltage v j (j = 2, ..., m) is expressed by Accordingly, the design of the multiple output switching power supply is facilitated. The circuit shown in Fig. 1 is compared with a prior art power supply having a plurality of magnetically coupled windings on a choke coil inserted in a secondary circuit of a transformer. Fig. 3 shows a configuration of the prior art power supply. T denotes a transformer, E denotes a D.C. input power supply, L denotes a choke coil having magnetically coupled windings n L1 and n L2 , and V₁ and V₂ denote voltages at output terminals 1 and 2, respectively. Voltages are induced, during an on period of the transistor Q, across the windings of the transformer T and the choke coils L, with dotted ends being positive. From Fig. 3, it is seen that the following relationships must be met during the on period of the transistor Q. where V nL1 and V nL2 are voltages induced across the windings n L1 and n L2 of the choke coil L, respectively. In the formulas (9) and (10), the affects by the leakage inductances and winding resistances of the transformer T and the choke coil L, and diode voltage drops are omitted for the purpose of simplification. The voltage V nL2 of the formula (10) is expressed as follows. From the formulas (9), (10) and (11), From the formula (12), the voltage V₂ at the output terminal 2 is expressed by Since the impedances of the transformer T and the choke coil and the diode voltage drops are omitted in the formula (13) as described above, the formula (13) corresponds to the formula (7) which represents the voltage V₂ in the circuit of Fig. 1. As seen from the comparison of the formulas (7) and (13), the output voltage V₂ of Fig. 1 is essentially determined by the turn ratio of the secondary windings n₁ and n₂ of the transformer T₁ while the voltage V₂ in the prior art of Fig. 3 cannot be defined without taking the voltage V L1 induced across the winding of the choke coil and the turn ratio of the choke coil L into consideration, and hence proper design of the power supply is difficult to attain. Referring to an equivalent circuit shown in Fig. 4, the operation of the prior art power supply of Fig. 3 is explained in detail. Fig. 4 shows the equivalent circuit during the on period of the transistor Q. E denotes the D.C. input power supply, I₁ and I₂ denote the currents supplied to loads connected to output terminals 1 and 2, respectively, Z₁ and Z₂ denote the impedances including the leakage inductances, winding resistances and circuit resistances of the transformer T, and Z L1 and Z L2 denote the impedances including the leakage inductances and winding resistances of the windings N L1 and n L2 of the choke coil L. As seen from Fig. 4, the currents I₁ and I₂ flowing through the output terminals 1 and 2 meet the following relationship. From the formula (14), the voltage V₂ at the output terminal 2 is expressed by From the formula (15), it is seen that when the output currents I₁ and I₂ change, voltage variations which are proportional to (Z₁ + Z L1 ) and (Z₂ + Z L2 ), respectively, take place in V₂. On the other hand, in the present case, it is seen from the formula (2) that the variation of V₂ when I₁ and I₂ change is proportional to Z₁ and Z₂. As a result, the voltage variation due to the change of the load current is smaller in the circuit of Fig. 1 and the precision of the output is higher. As described above, the circuits of Fig. 1 and of Fig. 3 are different from each other in their concepts and effects. Fig. 5 shows an embodiment of the invention with a specific circuit configuration of the energy discharge means 10 shown in Fig. 1. The energy discharge means 10 comprises windings n s1 and n s2 magnetically coupled to the primary winding n s of the reactor T₂ and diodes D₃ and D₄ connected to ends of the windings n s1 and n s2 , respectively. The winding n s1 is connected through the diode D₃ in parallel to the secondary winding n₁ of the transformer T₁ and the winding n s2 is connected through the diode D₄ in parallel to the winding n₂. During the on period of the transistor Q, voltages are induced across the windings of T₂ with the dotted ends being positive. Accordingly, the voltages induced across the windings n s1 and n s2 during the on period of the transistor Q are blocked by the diodes D₃ and D₄ and no current flows. The current flowing through the winding n s is an exciting current of T₂ and a magnetic energy is stored by the exciting current. When the transistor Q is turned off, the current flowing through the winding n s is cut off and voltages are developed across the windings of T₂ with the opposite ends to the dotted ends being positive, by the magnetic energy stored in T₂. As a result, currents flow out of the windings n s1 and n s2 into the loads connected to the output terminals 1 and 2, respectively. The operation of the energy discharge means is explained in detail with reference to an equivalent circuit of Fig. 6 during the off period of the transistor Q. L₂ denote an inductance of the reactor T₂, and Z₁' and Z₂' denote impedances including the leakage inductances, winding resistances and circuit resistances of the windings n s1 and n s2 , respectively. As seen from the equivalent circuit, the voltage V₂ at the output terminal 2 is expressed by V₂ = I₁·Z₁' + V D3 - (I₂·Z₂' + V D4 ) + V₁ (16) The voltages induced across the windings of T₂ during the off period of the transistor Q is expressed by the formula (4) as seen from Fig. 5, and the voltage V ns2 across the winding n s2 is expressed by From the equation (17), the voltage V₂ at the output terminal 2 is expressed by In the formulas (16) - (18), V D3 and V D4 are voltage drops across the respective diodes D₃ and D₄. The voltage drop I₁·Z₂' + V D3 at the output terminal 1 is expressed by k₁, and the voltage drop I₂·Z₂' + V D4 at the output terminal 2 is expressed by k₂. Thus, from the formula (18), V₂ is expressed by It is seen from the above that the voltage V₂ at the output terminal 2 during the off period of the transistor Q is defined by a function similar to that shown in the formula (6). The same is applicable to the multiple output switching power supply having m outputs. The output voltage V j (j = 2, 3, ..., m) is expressed by Accordingly, the output voltage V j is essentially defined by the turn ratio of the secondary windings n₁ and n j of the reactor T₂ as is done for the transformer T₁ and the design of the power supply is facilitated. Further, in the present embodiment, since the magnetic energy stored in the reactor T₂ is supplied to the loads by the energy discharge means 10, the efficiency of the multiple output switching power supply is improved. Fig. 7 shows another embodiment. In the present embodiment, a voltage V₁ at an output terminal 2 is a sum of a voltage V₁ developed across a winding n₁ and a voltage developed across a winding n₂. Windings n s1 and n s2 are similarly configured to develop a voltage V₂ at an output terminal 2. In the embodiment of Fig. 5, the turn ratio of the windings n₁ and n₂ is essentially same as the ratio of the output voltages V₁ and V₂, as shown in the formula (7). The same is true for the turn ratio of the windings n s1 and n s2 . On the other hand, it is seen from the formulas (5) and (8) that the smaller the ratios n₂/n₁ and n s2 /n s1 are, more the variation of the voltage V₂ due to the change of the output current I₁ is suppressed. Thus, in the present embodiment, the numbers of turns of the windings n₂ and n s2 are reduced to suppress the variation of the voltage V₂ and improve the output precision. In the circuit configuration of Fig. 7, the numbers of turns of the windings n₂ and n s2 are selected such that the voltage V₂ at the output terminal 2 is a sum of the voltage V₁ at the output terminal 1 and the voltages developed across the windings n₂ and n s2 . Thus, the turn ratios of the windings n₂ and n s2 are selected as follows. Accordingly, the turn ratios n₂/n₁ and n s2 /n s1 are smaller than those of the embodiment of Fig. 1, and hence the variation of the voltage V₂ at the output terminal 2 is suppressed and the precision of the output is improved. Fig. 8 shows another embodiment. In the present embodiment, a control circuit 11 is provided to control the switching of the transistor Q to keep the voltage V₁ at the output terminal 1 constant. It is seen from the formulas (6) and (19) that if the voltage V₁ at the output terminal 1 is constant, the voltage at the output terminal 2 can be stabilized within a variation of a voltage drop across an impedance in the circuit. In the present embodiment, since the voltage V₁ at the output terminal 1 is kept constant, the voltage V₂ at the output terminal 2 need not take the change of V₁ into consideration as shown in the formulas (6) and (19), and the output precision of the voltage V₂ is further improved. Alternatively, the control circuit 11 of the present embodiment may be provided in the embodiment of Fig. 5 to control the switching of the transistor Q in order to keep the voltage V₁ of the output terminal 1 constant. Fig. 9 shows another embodiment. In the present embodiment, m outputs are supplied to the loads in the embodiment of Fig. 8. By controlling the switching of the transistor Q by the control circuit 11 to keep the voltage V₁ at the output terminal 1 constant, m stable voltages can be supplied to the loads. Alternatively, the control circuit 11 may be provided in the embodiment of Fig. 5 to control the switching of the transistor Q in order to keep the voltage V₁ at the output terminal 1 constant, so that stable outputs are supplied from m secondary windings wound on the transfermer T₁ and the reactor T₂. The control circuit 11 shown in Figs. 8 and 9 may be Hitachi switching power supply control IC HA17524 or Texas Instruments TL 494. The configuration and operation of those IC's are well known and they are not explained here. In accordance with the present invention, the output voltages of the multiple output switching power supply can be defined by the turn ratios of the secondary windings of the transformer and the windings of the energy discharge circuit of the reactor, and the design of the power supply is facilitated. When the control circuit is used to control the switching device such that one of the output voltages is kept constant, the voltage variations of other outputs can be less than ±5%. Thus, the multiple output switching power supply used for the OA and FA equipments are simplified and the size and cost thereof can be reduced.

Claims: A multiple-output switching power supply comprising a switching device (Q), a transformer (T₁) for supplying current to a load during the ON period of the switching device (Q), and a reactor (T₂) for storing magnetic energy during the ON period and discharging the stored energy during the OFF period of the switching device (Q), wherein the primary windings (n p , n s ) of the transformer (T₁) and of the reactor (T₂) are connected in series with the switching device (Q) and a DC input power supply (E), wherein the secondary windings (n₁, n s1 ) each with a series-connected diode (D₁, D₃) are connected in parallel to said load, the diode (D₁) of the transformer (T₁) being conductive during the ON period and the diode (D₃) of the reactor (T₂) being conductive during the OFF period of the switching device (Q), wherein the ON-OFF switching of the switching device (Q) is controlled so as to keep the output voltage (V₁) supplied to said load constant, and wherein the reactor (T₂) comprises a further secondary winding (n s2 ) connected through a further diode (D₄) to a further load, said further diode (D₄) being conductive during the OFF period of the switching device (Q), characterized in that the transformer (T₁) comprises a further secondary winding (n₂) and a series-connected further diode (D₂), which further secondary winding (n₂) and diode (D₂) are connected in parallel to said further load, said further diode (D₂) of the transformer (T₁) being conductive during the ON period of the switching device (Q), and that the output voltage (V j ) supplied to said further load during said ON and OFF periods is given by the formula with n j = number of turns of said further secondary winding (n₂) of the transformer (T₂), k₁ = voltage drop in the secondary circuit comprising the secondary winding (n₁), kj = voltage drop in the secondary circuit comprising the further secondary winding (n s1 ). The power supply of claim 1, wherein the secondary windings (n₁, n₂; n₁, n m ) of the transformer (T₁) are connected in series and the secondary windings (n s1 , n s2 ; n s1 , n sm ) of the reactor (T₂) are connected in series.

Inventors: ABE, KOJIN, ONDA, KENICHI
Applicants: HITACHI, LTD.
Application year: 1985
Main classification: H02M3
IPC codes: H02M3