METHOD AND APPARATUS FOR CONTROLLING PWM INVERTER

Abstract: A method and an apparatus for producing a pulse width modulating signal to be used for operation of a PWM inverter wherein a modulating wave (U) of a sinusoidal waveform corresponding to the desired AC output voltage of the inverter is sampled at selected periodical time intervals (τ c ) to obtain a level of the modulating wave at each sampling point and the interval (Tu, Tv, Tw) is arithmetically deter­ mined between each sampling time point (A or C) and a time point (C or D) which appears between that sampling point and the next sampling point and at which time point a virtual triangular waveform (X) which has positive and negative peaks of a predetermined amplitude occurring alternately at the selected periodical time interval (τ c ), takes the same level as the sampled level of the modulating wave. On the basis of the time interval (Tu, Tv, Tw), the PWM signal (Eu, Ev, Ew) is produced.

Description: The present invention relates generally to a method and an apparatus for controlling a pulse width modulation (PWM) inverter which serves as an alternating current (AC) power supply source of variable voltage and frequency and, more particularly, to a method and an apparatus for producing a PWM signal for the PWM inverter control. A method and an apparatus according to the precharacterizing part of claims 1 and 6 respectively one known from IEEE PROC. VOL. 128, No.6 NOV.1981, PP 293-305. An asymetric PWM inverter process using a microprocessor is further known from "Preprints of the Third IFAC Symposium Lausanne", 12-14 September 1983, pp 453-459 Heretofore, the control apparatus for producing a PWM signal for controlling a PWM inverter is composed of a circuit for generating a modulating waveform such as a sinusoidal waveform or a stepped waveform approximating to the sinusoidal wave and having a frequency and an amplitude corresponding to those of the AC voltage to be produced by the PWM inverter, a carrier generating circuit for generating a carrier of triangular or saw-tooth waveform, and a PWM signal synthesizing circuit for deriving the PWM signal through comparison of the modulating wave with the carrier, as is described, for example, in the article "Intrinsic Harmonics of Idealized Inverter PWM System" by James W. A. Wilson and Jerome A. Yeamans, IAS 76 Annual, pp. 967-937. In the carrier generating circuit, the use of an oscillator capable of varying the output frequency thereof over a wide range is necessary for changing the frequency of the carrier in accordance with the frequency of the modulating signal with a view to enhancing the availability or utilization factor of the AC output voltage of the inverter on one hand and for increasing the frequency of the carrier for the purpose of improving the form factor of the AC output voltage on the other hand, resulting in a complicated circuit configuration and expensiveness of the control apparatus. It is an object of the present invention to provide a method and an apparatus for generating a pulse-width modulation (PWM) signal to be used for operating a pulse-width modulation (PWM) inverter system which serves as an alternating current (AC) power supply source of a variable voltage and frequency, said method and apparatus providing a good response characteristic for the generation of said pulse-width modulation (PWM) signal with high accuracy. In solution of the above object a method for generating a pulse width modulation signal according to the preamble of claim 1 is characterized by its method steps VI to VIII. An apparatus according to the preamble of claim 6 which solves the above object is provided by the characterizing part of claim 6. The subclaims 2 to 5 characterize advantageous developments of the method claim 1. The subclaims 7 to 10 characterize advantageous developments of the teaching characterized by claim 6. The invention will be described in more detail in conjunction with the accompanying drawings, in which: Figure 1 is a block diagram showing an arrangement of a control apparatus according to an embodiment of the present invention; Figure 2 is a view for illustrating the underlying concept of the invention for deriving the PWM signal when the amplitude ratio is smaller than 1 (unity); Figure 3 is a view for illustrating the principle of the invention for deriving the PWM signal when the amplitude ratio is greater than 1; Figure 4 is a view showing an arrangement of a counter circuit shown in Figure 1; Figure 5 is a view illustrating a process for determining the phase of the modulating wave at each sampling point with the aid of the counter circuit shown in Figure 4; and Figures 6A and 6B are flow charts showing processes executed by a micro-computer shown in Figure 1. Now, the invention will be described by referring to Figure 1 which shows schematically a general arrangement of the control apparatus according to an exemplary embodiment of the invention. A frequency command voltage f R indicative of the frequency of a sinusoidal modulating wave corresponding to that of the AC output voltage of an inverter system is generated by means of a potentiometer 10 and supplied to a voltage/frequency (V/F) converter 11. A series of pulses is produced by the V/F converter 11 with a period which is in proportion to the value of the frequency command voltage f R and applied to a counter circuit 14 at the clock terminal K as a clock signal. The content of the counter circuit 14 is incremented in response to the clock signal until the count value has attained a value corresponding to one cycle of the sinusoidal wave, whereupon a pulse T₁ is produced from the output Q of the counter circuit 14. This pulse signal T₁ is applied to a reset terminal R of the counter circuit 14, whereby the content thereof is reset. Subsequently, the counter circuit 14 starts again the counting in the incrementing direction in response to the clock pulses. This operation of the counter circuit 14 is periodically repeated. The content of the counter circuit 14 is read out by a single-chip micro-computer 13 at time points which correspond to sampling points of the sinusoidal wave. On the basis of the content of the counter circuit, phase positions ϑ of the sinusoidal wave at the sampling points are determined by the micro-computer 13, as will be described in detail hereinafter. The micro-computer 13 includes a group of memories such as RAM, EPROM, counter devices input/output ports and others. The micro-computer 13 fetches also a digital signal produced by an analogue-to-digital or A/D converter 12 which serves to convert a voltage command signal KH relating to the amplitude value of the aforementioned sinusoidal modulating wave into a digital signal. On the basis of the values of ϑ and KH, the micro-computer 13 arithmetically determines the pulse width of a pulse width reference signal in accordance with an expression (1) mentioned hereinafter, which pulse width reference signal in turn provides a basic quantity in determining the PWM signal. This pulse width is equal to a time interval (τ₋= AB or τ+= CD in Figure 2) between one sampling point and the time point at which the aforementioned virtual triangular wave takes the same level as that of the sinusoidal wave at the one sampling point. The pulse width thus determined is set at a programmable timer 15 constituted by a sort of counter. The micro-computer 13 generates a rectangular signal Es which takes a high level and a low level alternately every period τ c which corresponds to a half-cycle of the aforementioned virtual triangular wave, the signal Es being supplied to a latch circuit 16. The programmable timer 15 serves to produce a pulse width reference signal TU on the basis of the set value of the pulse width in the following manner. The value of τ₋ or τ+ calculated by the microcomputer is set into the programmable timer 15 in response to an interruption signal produced in synchronism with the leading or trailing edge of the signal Es. The programmable timer 15 is initiallized upon setting the value of τ₋ or τ+ so as to reset the content of its counter and change its output from low level to high level, and then starts to count an enable signal E of a constant frequency, as clock signal, derived from the microcomputer 13. When the counts of the programmable timer 15 reach the set value of τ₋ or τ+, its output is changed from the high level to the low level. Thus, the programmable timer 15 produces, as its output, the pulse width reference signal Tu having a pulse width of τ₋ or τ+. The PWM signal Eu is produced by a latch circuit 16 on the basis of the pulse width reference signal Tu and the signal Es in a manner as mentioned below. The latch circuit 16 is constituted by a D-type flip-flop having a data terminal supplied with the signal Es and a clock terminal supplied with the signal Tu. At the time point corresponding to the trailing edge of the signal Tu, the latch circuit 16 fetches the signal Es and makes the PWM signal Eu high level when the signal Es as fetched is at high level and low level when the signal fetched Es is at low level. In the case of a polyphase inverter, the programmable timer 15 produces a plurality of the pulse width reference signals different phases, respectively, corresponding to the phases of the inverter and, a number of latch circuits 16 are provided correspondingly to the pulse width reference signals, respectively, to thereby produce the PWM signal for each of the phases of the inverter. For example, in the case of the three-phase inverter system as illustrated, the programmable timer 15 is so arranged as to produce in addition to the signal Tu the pulse width reference signals Tv and Tw which are of the same waveform as the signal Tu but are out of phase with the latter by 120° and 240°, respectively, wherein the three-phase PWM signals Eu, Ev and Ew are produced by the latch circuits 16 provided in correspondence with the pulse width reference signal Tu, Tv and Tw. A suppression signal Sa supplied to the latch circuits 16 from the microcomputer 13 is utilized for suppressing the outputs of these latch circuits until they have been set in the normal state upon initialization for activation of the control apparatus. The arrangement of the control apparatus according to the invention has been outlined above. Now, the principle of modulation contemplated by the invention will be elucidated in more detail by referring to Figures 2 and 3. Figure 2 illustrates the principle of modulation when the amplitude ratio KH (i.e. ratio of the peak value E₁ of the sinusoidal waveform and the peak value H of the virtual triangular waveform is less than 1 (unity), while Figure 3 illustrates the modulation when the amplitude ratio is greater than 1 (unity). In both figures, the triangular waveforms are not actually produced but depicted for the convenience of description and thus referred to as the virtual waveform. In the following, the process for deriving the PWM signal Eu will be elucidated by using symbols shown in Figure 2. The micro-computer 13 is so programmed that interrupt request signals (hereinafter also referred to as IRQ signal) are produced at time points corresponding to the positive and negative peak points of the virtual triangular waveform X with a periodical time interval or period τ c corresponding to a half period of the virtual triangular waveform X which corresponds to the carrier in the hitherto known PWM inverter control apparatus. In response to the IRQ signal, the level A of the sinusoidal waveform U having a frequency determined by the frequency command f R and an amplitude E₁ determined by the voltage command K H =E₁/H (where H represents the amplitude of the virtual triangular wave which is constant) is determined in accordance with A=E₁ sinϑ at a time point when the IRQ signal is generated (e.g. for the phase position ϑ₁ corresponding to the first IRQ 1, the level A is given by A=E₁ sinϑ₁). Subsequently, the time interval or width τ₋ between the point A and a point B at which a portion of the virtual triangular wave between its peak corresponding to the IRQ 1 signal and its next peak of opposite polarity takes the same level as the level A is calculated in accordance with the following expression (1) to give the pulse width of the pulse width reference signal. τ₋= τ c 2 (1- E₁ H · sinϑ₁) = τ c 2 (1-K H · sinϑ₁) The value of the pulse width τ₋ as calculated in accordance with the above expression (1) differs slightly from the value obtained by the hitherto known modulation system (which value is derived from an intersection B' between the sinusoidal waveform represented by a broken curve and the triangular waveform). It has been, however, ascertained that such difference provides no problem in practical application by selecting the period or time interval τ c of the IRQ signal to be sufficiently short when compared with the frequency f₁ of the sinusoidal waveform (e.g. 1/τ c >5f₁). In response to the interrupt signal IRQ 1, the micro-computer calculates the value of τ₋ in accordance with the expression (1), and the value thus determined is set in the programmable timer 15 immediately before occurrence of the next interrupt signal IRQ 2. Then, the programmable timer 15 starts counting of the enable signal E, as clock pulse, supplied from the micro-computer 13, to thereby produce the pulse width reference signal Tu which is at high level from the start of the counting to a time point when the counted value reaches the set value mentioned above and rendered low level thereafter. In this way, when the interrupt signal IRQ 1 makes appearance, the micro-computer 13 calculates the value of the pulse width τ₋, which is followed by generation of the pulse width reference signal Tu by the programmable timer 15 at a time point substantially corresponding to the next interrupt signal IRQ 2, wherein the generated pulse width reference signal is initially at high level and takes low level after lapse of time equal to τ₋. For the purpose of facilitating the understanding of relationship between the pulse width τ₋ and the pulse width reference signal Tu, waveforms illustrated in Figure 2 are so plotted that the signal Tu corresponding to the value of τ₋ determined in response to the interrupt signal IRQ 1 assumes high level simultaneously with occurrence of the interrupt signal IRQ 1. It should however be appreciated that the signal Tu takes in reality the high level at the time point substantially corresponding to the next interrupt signal IRQ 2 with a delay almost equal to the period τ c of the interrupt signal IRQ. Similarly, the signal Es and the PWM signal Eu described below are in reality generated with the same delay as that of the signal Tu, although these signals are illustrated in Figure 2 on the assumption that they are generated without delay. Further, it should be appreciated that the following description is also made on the assumption that the delay is neglected for convenience sake. As described hereinbefore, the micro-computer 13 produces the rectangular signal Es having high level and low level alternating at every time interval τ c . This signal Es corresponds to the signal which assumes low level during a time interval corresponding to that of negative slope of the virtual triangular wave (i.e. time interval between the interrupt signal IRQ 1 and IRQ 2) while assuming high level during the time interval corresponding to that of the succeeding positive slope of the virtual triangular wave (i.e. time interval between the signals IRQ 2 and IRQ 3) in synchronism with the occurrence of the interrupt signal IRQ. Although the signal Es is so illustrated in Figure 2 that low and high levels thereof correspond, respectively, to negative and positive slopes of the virtual triangular waveforms, it should be appreciated that the correspondence may be reversed such that the signal Es are at high and low levels when the slope of the triangular waveform is positive and negative, respectively. In the latter case, the PWM signal described below is simply of the inverted form. According to the invention, the PWM signal is produced on the basis of the signals Es and Tu produced through the processes described above. More specifically, data or value of the signal Es (i.e. high level or low level) is fetched at a time point corresponding to the trailing edge of the signal Tu (the time point indicated by an arrow in Figure 2) to thereby produce the PWM signal. In more concrete, reference is again made to Figure 2. It will be seen that the rectangular signal Es is at low level during the time interval between the interrupt signals IRQ 1 and IRQ 2 in correspondence to the negative slope of the virtual triangular waveform. By fetching or sampling the rectangular signal Es of low level after elapse of time τ₋ at the time point corresponding to the trailing edge of the pulse width reference signal Tu outputted by the programmable timer 15, the PWM signal Eu transits from high level to low level which is held until the time point corresponding to the trailing edge of the next pulse width reference signal Tu outputted from the timer 15. Next, description will be made as to how the PWM signal is produced in the time interval between the interrupt signals IRQ 2 and IRQ 3. Also at the time point corresponding to the appearance of the signal IRQ 2, the pulse width τ₋ of the pulse width reference signal Tu to be set in the programmable timer 15 is determined through the procedure similar to that adopted in determining the pulse width τ₋ in response to the interrupt signal IRQ 1. More specifically, in response to the interrupt signal IRQ 2, phase position ϑ₂ of the sinusoidal wave is determined, which is followed by determination of the level C of the sinusoidal wave at a point C corresponding to the phase position ϑ₂ in accordance with C=E₁ sinϑ₂. Subsequently, the time interval τ+ between the point C and a point D which lies on the positive slope of the virtual triangular wave rising up in response to the signal IRQ 2 and at which the virtual triangular wave takes the same level as the level of the sinusoidal wave at the point C is calculated with the aid of the micro-computer 13 in accordance with the following expression: τ+= τ c 2 (1+K H · sinϑ₂) where K H =E₁/H. When the expressions (1) and (2) are compared with each other, it is seen that they differ in respect with the sign of the second term on the right hand side. Accordingly, both expressions (1) and (2) can be represented by a generalized form: τ= τ c 2 (1+K S · K H · sinϑ) where K S =1 when Es is at low level, and K S =-1 when Es is at high level. The pulse width τ+ of the pulse width reference signal Tu resulting from the arithmetic determination in accordance with the expression (2) is set at the program timer 15. The pulse width reference signal Tu outputted from the timer 5 through the process similar to that described hereinbefore assumes high level substantially in synchronism with the IRQ signal and returns low level after lapse of time τ+. The latch circuit 16 fetches the data of the signal Es (i.e. data of high or low level of the signal Es) at a time point corresponding to the trailing edge of the signal Tu falling to the low level, resulting in that the PWM signal Eu transits to the level corresponding to that of the signal Es, i.e. high level in the illustrated case. Succeeding to the time point corresponding to the interrupt signal IRQ 2, the processes as described are repeated thereby to produce the PWM signal Eu of the waveform as illustrated in Figure 2. The foregoing description has been made on the assumption that the rectangular waveform signal Es is sampled at the time point corresponding to the high-to-low level transition of the pulse width reference signal Tu to produce the PWM signal. However, it will be understood that same result can equally be obtained in case the pulse width reference signal Tu is inverted. In this case, the rectangular waveform signal Es is fetched or sampled at the leading edge or the low-to-high level transition of the inverted signal Tu . In any case, the rectangular waveform signal Es is fetched after the time lapse which corresponds to the value set at the programmable timer 15. Next, referring to Figure 3, description will be made concerning a method of increasing the voltage utilization factor by using the PWM system described above. In application where the PWM signal Eu is employed, the output voltage becomes at maximum when the amplitude ratio K H (=E₁/H) is 1 (unity). For increasing the voltage further, the amplitude ratio KH need to be increased higher than 1. Figure 3 shows a relationship between the sinusoidal waveform and the virtual triangular waveform for the amplitude ratio greater than 1. At the time points corresponding to the interrupt signals IRQ 4 and IRQ 5, the peak value H of the triangular wave is less than the peak value E₁ of the sinusoidal waveform. Thus, the amplitude ratio is greater than 1. Under the condition, a method of forming the PWM signal at these time points will be described. At the time point of appearance of the interrupt signal IRQ 4, the corresponding phase position ϑ₄ of the sinusoidal wave is fetched by the micro-computer 13 to determine the value of the pulse width τ+ of the pulse width reference signal Tu in accordance with the expression (2). Since the value of KH sinϑ₄ is greater than 1 at this time, the pulse width τ+ of the pulse width reference signal Tu is greater than the value of the period τ c of the IRQ signal. The value of the pulse width τ+ is set at the programmable timer 15. At a succeeding time point corresponding to the IRQ 5, the slope of the virtual triangular wave becomes negative, and the rectangular waveform Es transits to the low level. Accordingly, for determining the value of τ₋ in accordance with the expression (1), the phase position ϑ₅ at the time point corresponding to the interrupt signal IRQ 5 is read and the value of the term K H sin ϑ₅ is determined, the result of which will, however, show that the value of this term is greater than 1, and thus the value of τ₋ is negative. In this case, the value of τ₋ is set to zero. In other words, no value is loaded in the programmable timer 15, and thus the range of τ₋ is so determined as to satisfy the condition that τ₋≧0. Under the conditions, the data τ+ determined at the time point corresponding to the preceding interrupt signal IRQ 4 remains in the programmable timer 15. Accordingly, after the time lapse of τ+, the output signal Tu of the programmable timer 15 changes from the high level to the low level at a point G shown in Figure 3. In this case, since the value of τ+ is greater than τ c , the point G occurs during the low level state of the rectangular signal Es, involving no change in the sign of the PWM signal Eu, as shown in Figure 3, which means that the output voltage is correspondingly increased. When the number of such processes in which the sign of the PWM signal remains unchanged is increased, the output voltage is correspondingly increased. In this case, voltage regulation can be performed by varying the amplitude ratio KH. It will now be understood that the utilization factor of the output voltage can be increased through the simplified procedure. Next, a process for determining the level of the sinusoidal wave at every periodical time interval T c in response to the interrupt signal IRQ with the aid of the counter circuit 14 will be described by referring to Figures 4 and 5. The counter circuit 14 is provided with three counters A, B and C as shown in Figure 4. The counter A counts the pulses derived from the V/F converter 11 and produces at the terminal Q a pulse signal T₁ when its counts reach a value corresponding to one cycle ϑ'u of the sinusoidal wave U. The signal T₁ is applied to a reset terminal of the counter A thereby resetting the same. Since the signal T₁ is produced each one cycle of the sinusoidal wave U, the count value of the counter A represents the phase position in each cycle of the sinusoidal wave U. The count value of the counter A is fetched by the micro-computer 13 in response to each interrupt signal. The counter B also counts the pulses derived from the V/F converter 11 and produces a pulse signal T₂ when its counts reach a value corresponding to a predetermined phase angle Δϑ. The signal T₂ is applied to a reset terminal R of the counter B thereby resetting the same and also to the gate terminal G of the counter C. The period τ c of the interrupt signal IRQ may be either constant or variable so as to be inversely proportional to the frequency f R of the sinusoidal wave. In the latter case, the interrupt signal IRQ is produced periodically at a predetermined phase interval of the sinusoidal wave irrespective of its frequency. By selecting the value of Δϑ as above-mentioned equal to the last-mentioned predetermined phase interval, the pulse signal T₂ may be used as the interrupt signal in the mode of the period τ c being variable and the period τ c is given by the following equation: τ c = Δϑ 360° × 1 f R The counter C counts the enable signal 17, as clock pulse, supplied from the micro-computer 13 and the counting operation thereof is reset by the signal T₂ so that the content of the counter C just before resetting the same represents as time interval between two successive pulse signals T₂, that is the value of τ c in the variable τ c mode where the signal T₂ is used as the interrupt signal. The value of τ c obtained by the counter C is taken by the micro-computer to be used for calculation of τ₋ or τ+ in the variable τ c mode. In the mode of τ c being constant, the interrupt signal IRQ is produced by a PWM timer (not shown) contained in the micro-computer by setting into the timer data corresponding to the constant value of τ c so that the timer produces an output, as the interrupt signal, periodically at a time interval of τ c . In any case, the micro-computer 13 fetches the count value of the counter A in response to the interrupt signal. As mentioned hereinbefore, the count value of the counter A represents the phase position ϑ of the sinusoidal wave at the time point when the interrupt signal appears. The micro-computer 13 reads the value of sin ϑ out of a table which is provided in the micro-computer and stores the values of sin ϑ for various values of ϑ. The value of sin ϑ thus read-out is used for calculation of τ₋ and τ+. In the mode of the value of τ c being constant, the noise produced from a motor driven at a low speeds by the PWM inverter can be reduced, while in the mode of the value of τ c being variable, the balance between different phases is assured, which is especially advantageous when the frequency f R is high. Thus, the advantages of the respective modes can be realized by operating the apparatus in the constant τ c mode when the frequency f R is less than a predetermined frequency f o and in the variable τ c mode when the frequency f R is not less than the frequency f o . In this combined mode, the micro-computer 13 is programmed to compare the frequency command f R with the predetermined frequency f o and use the output of the PWM timer as the interrupt signal when f R <f o and the output signal T₂ of the counter B provided in the counter circuit 14 as the interrupt signal when f R ≧f o . Figure 5 shows a relationship between the count value of the counter A and the value sinϑ of the sinusoidal wave, typically in the phase U of a three-phase inverter. As mentioned hereinbefore, the counter A produces the pulse T₁ when its counts reach a value m corresponding to one cycle ϑ'u of the sinusoidal wave and the pulse T₁ resets the counter A. The count value n of the counter A is representative of the phase ϑu of the sinusoidal wave with a relationship therebetween given by an expression ϑu=n× 360° m . The micro-computer 13 fetches the value of n in response to the interrupt signal, calculates the value of ϑu according to the above expression and reads the value of sin ϑu out of the table contained therein. In the case of the three-phase inverter, the PWM signal has to be prepared for each of the phases U, V and W. In that case, the values of ϑ v and ϑ w of the V-phase and the W-phase can be determined in accordance with ϑ v =ϑ u -120° and ϑ w =ϑ u -240°, respectively, by making use of the value of ϑ u determined for the U-phase. The corresponding values of sin ϑ v and sin ϑ w are read out from the ROM. Next, processes executed by the micro-computer 13 for producing the three-phase PWM signals with the apparatus shown in Figure 1 will be described by referring to flow charts shown in Figures 6A and 6B. When the control apparatus is powered on, a main routine illustrated in Figure 6A is activated, whereupon initialization is effected at a step 601. This initialization process includes initialization of work areas of the memory, initialization of the program timer 15, setting of the mode of τ c , i.e. the constant τ c mode or the variable τ c mode or the combined mode, and the process for suppressing the generation of the PWM signal by producing the suppression signal Sa. At a next step 602, the frequency command f R is inputted and compared with a predetermined lower limit value (f₁) min of the frequency of the inverter. When the comparison results in that the frequency command f R is not greater than (f₁) min , the process proceeds to a step 603 where the suppression signal Sa is produced from the output port of the micro-computer 13 to thereby maintain the state in which the generator of the PWM signal is inhibited. When the commanded frequency f R exceeds (f₁) min , the process proceeds to a step 604 where the processing is executed for activating the PWM timer incorporated in the micro-computer. In the constant τ c mode, the counter or PWM timer of the micro-computer 13 is loaded with the data τ c which corresponds to a half of the period of the virtual triangular wave so that the IRQ signal shown in Figures 2 and 3 are produced. As described hereinbefore, the signal IRQ provides the interrupt signal for executing the PWM control processing. In the variable τ c mode, preparation is made such that output pulse T₂ of the counter B shown in Figure 4 is used as the IRQ signal. On the other hand, in the combined mode, this step includes comparison of the frequency f R with the predetermined frequency f o and determination as to which of the output of the PWM timer and the output T₂ of the counter B is used as the interrupt signal. Thus, the preparation procedure has been completed. Then, generation of the interrupt signal is awaited at a step 605. Upon appearance of the interrupt signal, execution of the interrupt routine illustrated in Figure 6B is started. At first, the level of the signal Es (which is set at high level at the stage of initialization) is inverted and the level resulting from the inversion is stored at the work area at a step 611. Next, at a step 612, the A/D converter 12 is activated to allow the command voltage K H to be inputted. At a step 613, the phase ϑ u of the U-phase sinusoidal wave is determined from the value available from the latch (Figure 4) of the counter circuit 14, and on the basis of the determined phase ϑ u the phase ϑ v of the V-phase sinusoidal wave is determined at a step 614. At a succeeding step 615, the values sin ϑ u and sin ϑ v of the sinusoidal waves which are corresponding to the phases positions ϑ u and ϑ v , respectively, are read out from the ROM. Next, at a step 616, the level of the signal Es (which is previously determined in correspondence with the positive or negative slope of the virtual triangular wave, e.g. high level for the positive slope and low level for the negative slope) is discriminated. In the case of the illustrated example, when the level of the signal Es is high, the process proceeds to a step 617 where the signs of sin ϑ u and sin ⁻ϑ v are inverted. On the other hand, when the signal Es is at low level, a dummy step is executed as indicated at a step 618. At the dummy step 618, the signs of sin ϑ u and sin ϑ v are left as they are. This dummy step is provided for the purpose of adjusting the processing time so that the succeeding step 619 may be executed after lapse of time equal to the time required for executing the step 617. Through the processing of the signs of the sin ϑ u and sin ϑ v , there can be eventually obtained the term K s sin ϑ u and K s sin ϑ v included in the expression (3). Next, at a step 619, value of sin ϑ w is arithmetically determined in accordance with sin ϑ w =(0-sin ϑ u -sin ϑ v ). Alternatively, it is of course possible to execute a step for determining ϑ w in succession to the step 614 to read out the value of sin ϑ w corresponding to the determined ϑ w from the table. On the basis of the data of sin ϑ u , sin ϑ v and sin ϑ w (inclusive of signs thereof) and the fetched data of K H , the values of Tu, Tv and Tw are arithmetically determined in accordance with the expression (3), i.e. Tu= τ c 2 (1+K S · K H · sin ϑ u ), Tv= τ c 2 (1+K S · K H · sin ϑ v ), and Tw= τ c 2 (1+K S · K H · sin ϑ w ) at a step 620. Next, at a step 621, it is decided whether the determined values Tu, Tv and Tw are positive or negative. When positive or 0 (zero), the determined values are set at the programmable timer 15 at a step 623, while for the negative values the values placed in the timer 15 until then are held as they are. Simultaneously, data of the signal Es (i.e. high level or low level) is set at the latch circuit 16. Subsequently, at a step 624, the suppression signal Sa is removed to enable the latch circuit 16. Further, it is confirmed at a step 625 whether or not the commanded frequency f R is greater than (f₁) min . When f R ≧(f₁) min , the step 605 of the main routine is restored for waiting for the next interrupt signal. When f R <(f₁) min , the step 603 of the main routine is regained to produce again the suppression signal Sa, to inhibit the output of the latch circuits 16 until f R ≧(f₁) min . When the data Es, Tu, Tv and Tw are set at the latch circuits and the programmable timer through the procedure described above, there are produced from these set values the signals for determining the levels of the PWM signals Eu, Ev and Ew by the latch circuits 16 in the succeeding interrupt cycle. As will be appreciated from the foregoing description, it is possible according to the invention to produce the PWM signal with a simple circuit configuration without necessity for providing the circuit for generating actually the carrier such as of triangular waveform. Further, the frequency of the virtual carrier can be varied in a facilitated manner through software technique. Besides, by virtue of the fact that the switching frequency which is optimum for the characteristics of power switching elements of the PWM inverter can be selected without modifying the configuration of the PWM control circuit. There is thus provided a PWM inverter which can enjoy an increased flexibility in use, to great advantages.

Claims: A method for generating a pulse width modulation (PWM) signal (Eu) to be used for operating a pulse width modulation (PWM) inverter comprising the steps of: (I) generating interrupt signals (IRQ₁, IRQ₂) periodically after selected periodical time intervals (τ c ), determining sampling time points (A, C); (II) generating a rectangular signal (Es) which assumes alternately high and low levels after each time interval (τ c ); (III) generating a modulating wave (U) whose level varies in the form of a sinusoidal waveform of a command frequency (f R ) and an amplitude (E1) determined by a given command amplitude ratio (K H ) to thereby determine the levels of the modulating wave (U) at the sampling time points (A, C), and sampling the modulating wave (U) after every time interval (τ c ); (IV) generating a virtual triangular wave (X) which has positive and negative peaks of a predetermined amplitude (H) corresponding in time to the sampling time points (A, C) and alternating at the sampling points (A, C) and (V) calculating time intervals of the pulse width modulation signal (Eu) based on the modulating wave (U) and the triangular wave (X) characterized by the following steps: (VI) calculating a first time interval (τ₋) between a first sampling time point (A) and a first point (B) of the virtual triangular wave (X) and a second time interval (τ+) between a second sampling time point (C) and a second point (D) of said virtual triangular wave (X), wherein said first and second sampling time points (A) and (C) are corresponding to successive peaks of one and the opposite polarity of said virtual triangular wave, respectively, said first point (B) is positioned on each segment of said virtual triangular wave extending between its peak corresponding to said first sampling time point (A) and its succeeding peak of the opposite polarity and at the same level as that of said modulating waveform at said first sampling time point (A), and said second point (D) is positioned on each segment of said virtual triangular wave extending between its peak corresponding to said second sampling time point (C) and its succeeding peak of the opposite polarity and at the same level as that of said modulating waveform at said second sampling time point (C); (VII) generating a pulse width reference signal (Tu) based on said first and second time intervals (τ₋, τ+), said pulse width reference signal being indicative of time points which correspond to time lapses of said first and second time intervals (τ₋, τ+) from said first and second sampling time points (A, C), respectively, and (VIII) generating the PWM signal (Eu) on the basis of said time points indicated by said pulse width reference signal (Tu) such that the level of said PWM signal is changed at each of said time points to a value corresponding to the level of said rectangular signal (Es) taken at that time point and held at said value until the next succeeding time point, wherein said pulse width reference signal (Tu) and said PWM signal (Eu) are generated with a delay almost equal to the period (τ c ) of the interrupt signal (at IRQ₂), while the first time interval (τ₋) is calculated during a first interrupt period starting at the former interrupt signal (IRQ₁). A method according to claim 1, wherein in said step III the phase angle (ϑ) of said sinusoidal wave at each of said sampling time points and a value of sin ϑ from said phase are determined, wherein said time interval (τ₋ or τ+) is arithmetically determined in accordance with expressions given by τ₋= τ c 2 (1-K H · sin ϑ) τ+= τ c 2 (1+K H · sin ϑ) where K H is a value given by said command voltage and representative of the ratio between the amplitude (E₁) of said sinusoidal wave and the amplitude (H) of said virtual triangular wave. A method according to claim 1, wherein said periodical time interval (τ c ) is constant. A method according to claim 1, wherein said periodical time interval (τ c ) is variable in dependence on said command frequency (f R ) so that the number of samplings performed during a single cycle of said modulating waveform is constant. A method according to claim 1, wherein said periodical time interval (τ c ) is constant when the value of said command frequency (f R ) is in a predetermined low frequency range, while said periodical time interval (τ c ) is variable in dependence on said command frequency (f R ) so that the number of the samplings taking place during a single cycle of said modulating waveform is constant. A control apparatus for generating a pulse width modulation (PWM) signal (Eu) to be used for operating a pulse width modulation (PWM) inverter, comprising means (13, 14) for generating interrupt signals (IRQ₁, IRQ₂) periodically after selected periodical time interval (τ c ), determining sampling time points (A, C); means (13, 611) for generating a rectangular signal (Es) which assumes alternately high and low levels after each time interval (τ c ); means (13, 613, 615, 619) for generating a modulating wave (U) whose level varies in the form of a sinusoidal waveform of a command frequency (f R ) and an amplitude (E1) determined by a given command amplitude ratio (K H ) to thereby determine the levels of the modulating wave (U) at the sampling time points (A, C), and sampling the modulating wave (U) after every time interval (τ c ); a microcomputer (13) for generating a virtual triangular wave (X) which has positive and negative peaks of a predetermined amplitude (H) corresponding in time to the sampling time points (A, C) and alternating at the sampling points (A, C) and means (13) for calculating time intervals of the pulse width modulation signal (Eu) based on the modulating wave (U) and the triangular wave (X), characterized in that said microcomputer (13) calculates a first time interval (τ₋) between a first sampling time point (A) and a first point (B) of the virtual triangular wave (X) and a second time interval (τ+) between a second sampling time point (C) and a second point (D) of said virtual triangular wave (X), wherein said first and second sampling time points (A) and (C) are corresponding to successive peaks of one and the opposite polarity of said virtual triangular wave, respectively, said first point (B) is positioned on each segment of said virtual triangular wave extending between its peak corresponding to said first sampling time point (A) and its succeeding peak of the opposite polarity and at the same level as that of said modulating waveform at said first sampling time point (A), and said second point (D) is positioned on each segment of said virtual triangular wave extending between its peak corresponding to said second sampling time point (C) and its succeeding peak of the opposite polarity and at the same level as that of said modulating waveform at said second sampling time point (C); a timer (15) is provided for generating a pulse width reference signal (Tu) based on said first and second time intervals (τ₋, τ+), said pulse width reference signal being indicative of time points which correspond to time lapses of said first and second time intervals (τ₋, τ+) from said first and second sampling time points (A, C), respectively; and means (13-16) are provided for generating the PWM signal (Eu) on the basis of said time points indicated by said pulse width reference signal (Tu) such that the level of said PWM signal is changed at each of said time points to a value corresponding to the level of said rectangular signal (Es) taken at that time point and held at said value until the next succeeding time point, wherein said pulse width reference signal (Tu) and said PWM signal (Eu) are generated with a delay almost equal to the period (τ c ) of the interrupt signal (at IRQ₂), while the first time interval (τ₋) is calculated during a first interrupt period starting at the former interrupt signal (IRQ₁). A control apparatus according to claim 6, wherein said means (615; 619) for determining the level of said modulating wave include means (16) for determining the phase angle (ϑ) of said sinusoidal wave at each of said sampling time points and means for determining a value of sin ϑ from said phase angle, wherein said time interval (τ₋ or τ+) is arithmetically determined in accordance with expressions given by: τ₋= τ c 2 (1-K H · sin ϑ) τ+= τ c 2 (1+K H · sin ϑ) where K H is a value given by said command voltage and representative of the ratio between the amplitude (E₁) of said sinusoidal wave and the amplitude (H) of said virtual triangular wave. A control apparatus according to claim 6, wherein said periodical time interval (τ c ) is constant. A control apparatus according to claim 6, wherein said periodical time interval (τ c ) is variable in dependence on said command frequency (f R ) so that the number of samplings performed during a single cycle of said modulating waveform is constant. A control apparatus according to claim 6, wherein said periodical time interval (τ c ) is constant when the value of said command frequency (f R ) is in a predetermined low frequency range, while said periodical time interval (τ c ) is variable in dependence on said command frequency (f R ) so that the number of the samplings taking place during a single cycle of said modulating waveform is constant.

Inventors: HATTORI, MOTONOBU, MUTOH, NOBUYOSHI, NAGATO, YUICHIRO, SAKAI, KEIJIRO
Applicants: HITACHI, LTD.
Application year: 1984
Main classification: H02M7
IPC codes: H02M7