CIRCUIT FOR REDUCING ERRORS IN A DATA RECEIVER

Abstract: In a data receiver, a complex data signal after demodulation and sampling is applied to an equalizer (15) which provides an improved signal xn which is applied to a phase and amplitude correction circuit (17) for correction utilizing a complex reference vector (vn). The corrected signal is applied to a decision circuit (19) the input and output of which are applied to a difference determinator circuit (21), the output of which is a complex residual error signal (dn) utilized in conjunction with a plurality of gain factors ( gamma 0, gamma 1, delta 0, delta 1) generated in a gain factors generator (24) to determine the reference vector (vn). The gain factors ( gamma 0, gamma 1, delta 0, delta 1) vary in time from transmission initialization such that optimal compensation for transmission disturbances is achieved both at initialization and after stabilization.

Description: _ L _ CIRCUIT FOR REDUCING ERRORS IN A DATA RECEIVER Technical Field This invention relates to data receivers of the kind including circuit means compensating for transmis- sion disturbances in a carrier-modulated transmission system, which circuit means includes a correction circuit for correcting a previously equalized received signal, a decision circuit responsive to the corrected received signal to provide output data signals, a difference determinator circuit coupled to an input and an output of said decision circuit and a correction factor deter¬ minator circuit coupled to an output of said difference determinator circuit and an input of said correction circuit. Background Art Phase shift keying (PSK) - in particular differential PSK (DPSK) - and quadrature amplitude modulation (QAM) are presently used in many systems for data transmission. To represent n discrete data values, k different phase values are used by which a carrier signal is modulated at given clock times. The carrier signal is modulated to m different amplitude levels (for PSK: m=l) where n k.m. The data values can also be represented by complex values distributed over m circles, with the zero point as the center. These complex values are then modulated by a complex carrier. Finally only the real part of the modulated carrier is transmitted, usually after filtering. A representation using complex signals for transmission systems with carrier modulation leads to a brief and clear mathematical description. In a technical implementation, however, pairs of real signals are used instead of complex signals. Preferably these real signals should be related to the complex signals according to the complex number calculus. Assuming error free transmission, the discrete values transmitted would arrive at the receiver after OM < wπ demodulation of two carrier waves, shifted 90° with re¬ spect to each other, by which a complex baseband signal is generated, at sampling instants which were suitably derived. Since disturbances such as: signal distortion (intersymbol interference), additive noise, frequency shift, phase jitter, phase hits, gain variation, always occur during transmission, a- suitable means for error compensation must be provided in the receiver. It is practice in the art, to use equalizers for eliminating signal distortion (intersymbol interference). Some of these equalizers can adapt their characteristics to varying distortion conditions. A data receiver of the kind specified is known from U. S. Patent No. 4,091,331. In the known system the difference determinator provides an output repre¬ senting the phase difference between the phases of the input and output signals of the decision circuit. This phase difference is fed to a phase error predictor cir¬ cuit which includes a feedback path and which is con- trolled by a gain control signal adaptively set in accordance with the ratio between phase jitter and additive noise. Although the known circuit does result in improved performance where the ratio between phase jitter and additive noise varies, the known circuit has the disadvantage that an undesirable high number of er¬ rors does still occur. Disclosure of the Invention It is an object of the present invention to provide a data receiver of the kind specified wherein the aforementioned disadvantage is alleviated. Therefore, according to the present invention, there is provided a data receiver of the kind specified, characterized in that the difference determinator circuit is responsive to complex signals representing the corrected received signals and the output data signals to provide complex residual error signals, and in that the correction factor determinator circuit is responsive to the complex residual error signals and to a gain control signal varying in a time dependent manner from an initial signal value when the receiver initially starts to receive transmitted data to a final signal value different from the initial value, whereby the correction factor determinator circuit provides a com¬ plex reference signal applied in operation to the correction circuit for compensating for phase and amplitude errors in the data receiver. It will be appreciated that in a circuit according to the invention, since the gain control signal varies in accordance with the time state of the receive process an improved compensation for transmission disturbances is obtained as compared with the known system, in that the occurrence of errors is reduced, since good compensation for errors can be achieved both at initialization and after stabilization. Brief Description of the Drawings The invention will now be described by way of example with reference to the accompanying drawings, in which: Fig. 1 is a vector diagram illustrating the discrete states and errors that occur in phas.e shift keying modulation with eight phases (8-PSK); Fig. 2 is a vector diagram illustrating the discrete states and errors that occur in quadrature amplitude modulation with eight target points (8-QAM); Fig. 3 is a block diagram of a receiver (with- out demodulator stage) for recovering discrete data values which were transmitted by modulating the phase and amplitude of a carrier at discrete times;: Fig. 4 illustrates, in a signal flow diagram, the details of a circuit arrangement, helpful in under- standing the present invention, that consists of a single proportional feedback loop; Fig. 5 illustrates, in a signal flow diagram, the details of a circuit arrangement, helpful in under¬ standing the present invention, that consists of a proportional and differential feedback loop; Fig. 6 illustrates, in a signal diagram, the details of the reference vector determinator included in the circuit of Fig. 3; Fig. 7 illustrates another embodiment of the reference vector determinator included in the circuit of Fig. 3; Fig. 8 illustrates, in a signal diagram, the details of the gain factors generator included in the circuit of Fig. 3; Fig. 9 illustrates, in graphical form, an example of the introduction of phase errors caused by- frequency shift and phase jitter, and the compensation of the phase, error in case of frequency shift; and Figs. 10 and 11 illustrate the time variation of the gain factors generated by the circuit of Fig. 8. Best Mode for Carrying Out the Invention Referring first to Fig. 1, there is shown a signal diagram representing the eight possible discrete states of the signal parameter in an eight-phase PSK transmission system. Each of the eight different states, designated 1 to 8 in Fig. 1 represents a different three-bit code word. Each state corresponds to a com¬ plex value, or "pulse amplitude", which can be repre¬ sented either as an = exp(jώn) or as an„ = a where is termed the in-phase component and the quadrature component. Fig..2 shows, in a similar way to Fig. 1, a signal diagram representing the eight possible discrete states of the signal parameter in a 2-amplitude, 8-phase quadrature amplitude modulation (QAM) transmission sys- ZS<£?v tem. The comp c lex value an = cn.exp r H ι -"ώn) y , where cn is the amplitude. The system is considered time-discrete, i.e. the discrete values of the signal parameters are defined only for given "sampling times", that occur with period T. In a transmitter, a carrier signal, e.g. a sinusoidal wave, is modulated in a known manner at each sampling time by one of the eight discrete complex pulse amplitudes a (as shown in Figs. 1 and 2). This may be effected by modulating a complex carrier exp (jw t) in phase and amplitude by a complex value c .exp(j ) and transmitting the real part thereof: Re Cc n .exp(j(^ n ) .exp(jw c t)] Fig. 3 shows a receiver in which a demodulated complex signal z (in-phase and quadrature components) is sampled by a sampler 11 at intervals T, the sampling times being determined by a clock recovery circuit 13 which derives clock pulses from the received signal. The clock recovery circuit 13 also provides clock sig¬ nals CL for other units. From the sample values z thus obtained, the received .pulse amplitude must be d n eter¬ mined for recovering the transmitted information. A decision circuit 19 is provided for this purpose. How- ever, during transmission the signal is subject to sev¬ eral disturbances, depending on the characteristics of the transmission medium, such as signal distortions, additive noise, phase jitter, phase hits and small gain hits. Signal distortion is compensated for by an equalizer 15, which operates in discrete time, to obtain an improved signal x as shown in Fig. 1 or 2. However, this signal is still disturbed by phase jitter, additive noise and hits which must be reduced to the largest possible extent to allow correct pulse amplitude determination. To allow phase tracking, a phase and amplitude correction device 17, is provided, that will reduce phase errors and amplitude errors by multiplication with the complex conjugate of a refer- ence vector vn. The resulting 3 value still has residual phase errors and amplitude errors. A reference vector determinator 23, generates a new reference vec¬ tor v . from the previous reference vector v and the estimated in-phase and quadrature error components, in dependence on gain factors derived from a gain factors generating circuit 24, as will be more fully described hereinafter. These estimated errors are derived from the residual errors in a difference determinator circuit 21. A decision circuit 19 determines the output value a from y by selecting, from one of the eight discrete pulse amplitudes, the one which is closest to y . The difference determinator for the in-phase and w. quadrature component 21, receives both y and "a and furnishes the value of the residual error dn with an in-phase and a quadrature component, as input to the reference vector determinator 23. In good approxi¬ 2. a* stands for the complex conjugate of a ) The circuit illustrated in Fig. 4 is explan¬ atory of one aspect of an implementation for determin- ing the reference vector. Referring to Fig. 4, a feed¬ back path for obtaining a revised value v for the reference vector includes a multiplier 35, in which the residual error signal from the circuit 21 is multiplied by a given factor o . The weighted residual error Q .d is fed to a second multiplier 29, where the weighted residual error is multiplied with the reference vector iRBΛ OMPI &£ v from delay element 27 . This delay element 27 has a delay time of one sampling period T. The result of this last multip c lication vn . ^} .dn is added in an adder 31 to the value of the reference vector. The arrangement consisting of elements 35, 27, 29 and 31 provides a new value of the reference vector according to the e ,q"uation vn+,l.=vn + vn. rf.dn. (Note that the quantity Q is a real value in contrast to the complex value quantities vn and dn. ) The circuit illustrated in Fig. 5, also explanatory of certain aspects of an implementation for determining the reference vector, illustrates in addition to components corresponding to those in Fig. 4, a differential feedback loop the function of which is to compensate for frequency shift. Thus, the circuit of Fig. 5 additionally includes a multiplier 55, a delay element 51, and adders 57 and 37. The residual error d is weighted by multiplication with factor <fin 55, the result is added in the adder 57 to the contents of the delay element 51. The result of the addition in the adder 57, en, is an integrated residual error Σ dn. This value is added in the adder 37 to the weighted residual error o .6. . (Note that the quantity <Tis a real value.) Referring now to Fig. 6, which shows an imple¬ mentation of the reference vector determinator 23, double lines are used for complex value quantities and single lines for real or imaginary value quantities. It will be appreciated that, in a practical embodiment, complex valued quantities are represented by pairs of real valued quantities derived during the demodulation operation and real valued quantities are represented by digital signals. Furthermore, it should be understood that the various digital processing circuits such as adders, multipliers and delays may be implemented by a digital signal processor integrated circuit. It is assumed that the disturbances still present in x (see OMPI Fig. 3), i.e. after equalization are: noise, frequency shift, phase jitter, changes in nominal amplitude (small gain hits), and phase hits. Changes in amplitude can be introduced by several reasons, e.g. by impulse noise. The real part (in-phase component) of d is multiplied in the multiplier 35 with a factor O, . This provides one input to the proportional feedback circuit for amplitude correction. The result of this multiplication forms the real part of a complex value g , the output from an adder 47 forming the imaginary part of g . The complex value g constituted by its real part from the multiplier 35 and imaginary part from the adder 47, is multiplied in the multiplier 29 with the output signal from the delay element 27. The processing of the imaginary part up to the adder 47 is realized in a similar way to that shown for the complex values in Fig. 5 (up to adder 37), except for the provision of a multiplier 53 which is coupled to the output of the delay element 51. The factor 6 Q applied to the multiplier 53 controls the decrease in contribution of the contents of the delay element 51, e , to the addition in 57. (Note that e and O , O 2 ' <£ ' τ» £ Q are real valued quantities with 0 < J * . < 1, 0 < < 1, 0 < O, < 1, 0 < S 0 < 1) • It will be appreciated that e approximates a value that minimizes ' the averaged value of the quadrature component (imaginary part) in the residual error d . This value e would correspond to an averaged phase error in x caused by frequency shift. (Note that for small cC, sin & ^ c ^Stan cC ). Another embodiment of the reference vector determinator 23 will now be described with reference to Fig. 7. In Fig. 6 the quadrature component (imaginary part) of d was multiplied by a factor o in the multi- plier 45. In Fig. 7 a contribution q is added to 2f i n an adder 73. This contribution σ is derived from the quadrature component of d . The instantaneous corre- lation present in the quadrature component of d , p is derived by multiplication in a multiplier 63 of the quadrature component of d and its previous value, pro¬ vided by a delay element 61. The averaged correlation q is derived from p after multiplication in a multi- plier 65 by a factor and addition in an adder 71. The previous value of σ , provided by a delay element 67, is multiplied in a multiplier 69 by a factor . The result of this multiplication and that from the multiplier 65 are added in an adder 71. The value of q is a measure of the averaged correlation in the quad¬ rature component of d . (Note p , q , ζ, , and are real valued quantities with 0 < < 1, 0 < / < 1). The gain factors generator circuit 24 illus- trated in Fig. 8 generates the factors l, ,ni O -2,,n , 0 ι > -.• ( c\ « • These factors have initial values Q * Λ n , 0 ' cf " i o' O Q o' tϊιe values stabilizing finally at * ' ° i ' & 0 ' respectively. It should be under¬ stood that these final values are optimal under stable circumstances. The circuit 24 includes a feedback loop including a delay element 75 having an output coupled to a multiplier 77, an output of which is coupled to an adder 79 having an output fed back to an input of the delay element 75. The output of the adder 79 forms an output of the feedback loop for providing the factor from which are provided the gain factors ,n and n 0,n as will now be described. The content of the delay element 75 is set initially to 2>. Q over an input line 26. Multiplication by a factor t in the multiplier 77, and addition of r., in the adder 79 results after the first time interval in O . .. Thereafter, the delay element 75 behaves as a normal delay element, and 0 . ,„n is produced from* ~the out- put of the adder 79. A multiplier 81 produces from O "V, and the factor r 2 - An adder 83 produces f, from the output of the multiplier 82 and the constant r . . An adder 85 produces 5 0 n from the output of the multiplier 84 and constant r_ . (Note O , , Q ~ , <^l, ' ^ " θ , n nd r 0 ' r l' r 2' r 3 ' r 4' r 5 ' 6 are re l value quantities) . The time variation of the factor an d *y, i ' ϊl ^ with respect to the number of time intervals T is shown in Fig. 10 and the time variation of the fac¬ tors O n 0,n and θ1 ι ,„n with resp r ect to the number of time intervals T is shown in Fig. 11. In an environment where the receiver is con¬ nected to multiple transmitters each having their own transmission path, the behavior of the receiver imme¬ diately after the start of receiving requires special attention. Fast and proper initialization, followed by a fast stabilization of the circuit shown in Figs. 6 or 7, will save inactive time in the beginning of a new transmission period. For short transmission periods, with receiver input from different transmission paths, this behavior is of special interest. In the gain factors factors are initially set at and will finally converge to latter being their optimal values after stabilization. The values 0 . , « 2 ' ® > & Q should be optimal for the type of modualtion and the signalling rate that is used and the disturbances that can be expected in x , i.e., after equalization. Since the integration loop in Figs. 6 and 7, consisting of the elements 51, 53, 57 and 55 is intended to compensate for frequency shift, O Λ should be close to 0 and 0 _ close to 1. It will be appreciated that the values of o, and 0 determine the speed and gain of the integration loop. Initially, the content of the delay element 51, Figs. 6 and 7, is set to 0, and should converge to a quantity corresponding to the phase offset during one sampling period T introduced by frequency shift. The content e of the delay element 51 is derived by the following equations : e Q = 0 e n = θ ι l , n . Im( d n + n u , n n. e n n— Λ l n>l For accurate frequency shift compensation in the presence of disturbances other than frequency shift (like noise and phase jitter) should be close to 0 and should be even closer to 1. In the period shortly after initialization when n is a small number, a small value for o l., n will cause a slow stabilization of en , a hig 3 h value for o 1 ■> , „n will introduce a contribution from phase jitter and noise in en. In the c present embodiment a O1, ,„n decreases from a relatively high value to a low value. The distur- bance in e , introduced by phase jitter and noise, during the intervals after start up, can be diminished by - choosing Q 0,n to be less than 1. However, when O Q n differs from 1 the frequency shift compensation is reduced. Therefore o Q is caused to increase to 1. The decrease of ό. 1,n and the increase of ύ 0 n ,n should be optimal for the fast stabilization of frequency shift compensation in the presence of noise and phase jitter during transmission. The gain factors O 1,n and O _,_n are initially at higher values than their optimal values after stabilization, namely and . Furthermore, to diminish the negative effect of the contribution of α (see Fig. 7) in the periods shortly after initialization, bis selected to be very small, whereby σ will therefore increase slowly to its final value in the stabilized state. The contri¬ bution of σ in the periods shortly after initialization is not significant. The arrangement illustrated in Fig. 8 to derive the gain factors O l.,n , cT_2,n. ά,l,n„ and o n υ,„n can be described by the following equations: n-1 =r, ') + r r 1,0 j=l ^l,n =r 3" ^l,n + r 4 <? * 0,n =r 5' ^l,n + r 6 Since 0 < r n 0 < 1 the factors Q 1,n, # v 2,,,n. ύ, . ) n _ will converge to respectively o . Υ _ ^l,n = ^ = r l ^0 ^2,n ^ 2 - r 2' r l ^O *l.n 1 " r 3° r l + r 4 χ - r ι Fig. 9 shows an example of frequency shift and phase jitter disturbances present in x , i.e., after equalization. The x-axis in Fig. 9 represents the number of intervals T. In this example a frequency shift of 7Hz, and sinusoidal phase jitter of 10° peak-to-peak and 180 Hz repetition is assumed. In the receiver with a 1600 Baud signalling rate, 7Hz frequency shift corres- ponds to 1.575° phase change per sampling period T (or symbol interval), λ . . In Fig. 9, the phase jitter corresponds to phase changes between -1.6° and +1.6° per sampling period T. In this Fig. ^ φ . and Δ φ represent respectively the ideal and realized phase error compensation. The alternation in l 6 with respect to I AΛ φ . decreased in correspondenc r e , with a decrease in & 1, . The differences between Δ φ*r,n and Δ φ . have to be diminished because this kind of phase difference, introduced by phase jitter, results in lagging. In practice the phase of x will be disturbed also by other effects such as noise. Experimental results for an 4800 bps QAM modem receiver in the presence of various disturbances led to a set of favorable factor values. These disturbances were: - noise, C-message weighted, SNR (signal-to-noise ratio) 12-40 dB - frequency shift. 0-10Hz - phase jitter, 0-10° with repetition 50-200HZ and the factor values: *ι,o = 0 ' 31 * . . . 18 *2,0 - °- 31 2 = 18 £ 1/0 = 0.044 ; ^ = 0, ,002 ξ Q 0 = 0.958 ; £ Q = 0.99994 (It should be understood that C-message noise is the total frequency weighted noise measured in the absence of a signal) . During these experiments q (see Fig. 7) was kept to 0 by selecting the factor = °« These experi¬ mental results were concerned with the probability of ΓREΛI OMPI an error in the first 80 or 800 bits after the receiver training and with the probability of bit errors after stabilization. The decrease in O δ and O, , and the increase in 0 Q were realized by selecting suit¬ able values for r Q , r,, ..., r g (see Fig. 8; 0 < r. < 1 for i=l, 2, 3, 6 and -1 < r < 0, if i=4, 5). The experimental results were most favorable for a decrease in O l,n , O2,„n, l,,n„ and increase in . Q 0 realized by a factor r Q = 0.9946 and addition constant r. = 0.00098. Using the same 4800 bps modem as above, exper¬ iments were made where the contribution of σ was used. Most favorable factor values were: ° 1 , Q = °' 31 ' " ^1 = ° *15 ^2,0 = °' 27 * ^2 - ° ' 11 s = 0.044 1,0 - 0. 001 £ Q = 0.956 S Q = 99994 £ = 0.0625 ; / = 0.99976 The allowable SNR LEVEL (only C-message noise was considered) for a bit error rate of 10~ was 15.7 dB with the above set of factor values, and 16.1 dB with the set of constants for the situation where σ was kept 0 because = 0* However, in the presence of hits, the last set of factor values delivers an inferior per¬ formance compared with those of the first experiment. Initially the reference vector v is set to a value During an initialization that is part of a se r lf-recovery operation, the factors Ol,,n,' and O 1 - are advantageously set to larger values, the &£&> procedure for initializing the reference vector v is the same as above and the manner of changing the fac¬ tors al.,n. 0 ~2,n„ and O ,l,„n is maintained. When there exist large errors during detection, the error d may exceed a certain threshold. When- is occurs the refer¬ ence vector is initialized again, keeping the instan¬ taneous compensation e stored in the delay element 51 unchanged and also keeping the equalizer 15 unchanged. Under very bad conditions during transmission, it can be hard to train the receiver according to the principles of self-recovery equalization. Saving the compensation e that was built up during a self-recovery operation, and initializing the reference vector v again is advan- tageous in such conditions. Experiments for a 4800 bps QAM modem in the presence of much disturbance resulted in favorable results with the following values for l,n' °2,n' °l,n' <f, 0,n and: ^2,0 - °' 4 ; ^2 - °' 18 1 0 = 0.044 ; £ ± = 0 .002 0 0 = ° - 958 ' * o = ° - 99994 * £

Claims: CLAIMS : 1. A data receiver, including circuit means compensating for transmission disturbances in a carrier- modulated transmission system, which circuit means in¬ cludes a correction circuit (17) for correcting a pre- viously equalized received signal (x ), a decision cir¬ cuit (19) responsive to the corrected received signal (y ) to provide output data signals (a )» a difference determinator circuit (21) coupled to an input and an output of said decision circuit (19) and a correction factor determinator circuit (23) coupled to an output of said difference determinator circuit (21) and an input of said correction circuit (17), characterized in that said difference determinator circuit (21) is respon¬ sive to complex signals representing said corrected received signals (y ) and said output data signals (^ n ) to provide complex residual error signals (d ) , and in that said correction factor determinator circuit (23) is responsive to said complex residual error signals (d ) and to a gain control signal (<C# σ , 6 Q , ό, ) varying in a time dependent manner from an initial signal value when the receiver initially starts to receive transmitted data to a final signal value different from the initial value, whereby said correction factor determinator cir¬ cuit (23) provides a complex reference signal (v ) applied in operation to said correction circuit (17) for com¬ pensating for phase and amplitude errors in said data receiver. 2. A data receiver according to claim 1, characterized in that said correction factor determinator circuit (23) includes: a first compensation circuit (45, 47, 51, 53, 55, 57) responsive to the quadrature com¬ ponents of said complex residual error signals (d ) and to said gain control signal ( O , €> „ , Q f ) to provide a compensated output signal which is compensated for frequency shift; and a second compensation circuit (27, 29, 31, 35) responsive to the in-phase components of said complex residual error signals (d ), to said gain control signal ( 0 - , , 0 2 > 0' 1^ nd to sa "- d compensated output signal and adapted to provide said complex refer¬ ence signal (v ) . 3. A data receiver according to claim 2, characterized in that said gain control signal repre¬ sents a plurality of gain control factors (<3T, O ' , α n , »,) and in that said first compensation circuit includes a first adder circuit (47) having first and second inputs, a delay element (51) having an input coupled to said first input and an output coupled to a first multiplier circuit (53) for multiplication by a first one of said gain control factors ( ό Q ) , an output of said first multiplier circuit being coupled to a first input of a second adder circuit (57) having a second input coupled to an output of a second multiplier circuit (55) adapted to multiply the quadrature component of the complex residual error signal (d ) by a second one of said gain control factors ( o, ), an output of said second adder circuit (57) being coupled to an input of said delay element (51) and to the first input of said first adder circuit (47), the second input of said first adder cir¬ cuit (47) being coupled to an output of a third multi- plier circuit (45) adapted to multiply the quadrature component of the complex residual error signal (d ) by a third one of said gain control factors ( ° V ~.) . n 2 4. A data receiver according to claim 3, characterized in that said second compensation circuit includes: a fourth multiplier circuit (29) having a first complex input adapted to receive an output of said first adder circuit (47) as quadrature component and to receive, as in-phase component, an output of a fifth multiplier circuit (35) adapted to multiply the in-phase component of said complex residual error signal (d ) by a fourth one of said gain control factors ( a . ) , said fourth multiplier circuit (29) having a second complex input coupled to an output of a further delay element (27); a third adder circuit (31) having inputs coupled to an output of said fourth multiplier circuit (29) and said output of said further delay element (27) and an output coupled to an input of said further delay element (27) and adapted to provide said complex reference signal (v ) . 5. A data receiver according to claim 4, characterized by correlation circuit means (61, 63, 65, 67, 69, 71) responsive to the quadrature component of said complex residual error signal (d ) and adapted to provide averaged correlation signals (q ) arranged to modify said third one of said gain control factors ( 0 ~ ) , said third multiplier circuit (45) being arranged to multiply the quadrature component of the complex residual error signal (d ) by the modified third one of said gain control factors. 6. A data receiver according to claim 5, characterized by a fourth adder circuit (73) having in¬ puts adapted to receive said averaged correlation signal (g and said third one of said gain control factors ( f 2 ) anc ^ having an output coupled to an input of said third multiplier circuit (45). 7. A data receiver according to claim 6, characterized by a gain factors generator circuit (24) adapted to generate said first, second, third and fourth gain control factors ( ^ ι r o 2 > ^o' ^\) » said gain factors generator circuit including a feedback loop (75, 77, 79) and being responsive to a plurality of predetermined factors (r o~ r g) whereby each of said gain control factors ( f, , 6> 2 , <£ " , -, ) is dependent on an output of said feedback loop (75, 77, 79) and on at least one of said predetermined factors (r Q -r 6 ) . (Mϊ OMPI_ ^ "SVΠFO 8. A data receiver according to claim 7, characterized in that said first one of said gain con¬ trol factors ( ό Q ) is arranged to increase towards a value which is close to unity and said second one of said gain control factors (σ,) is arranged to decrease towards a value which is close to zero.

Inventors: KAMERMAN ADRIAAN, KAMERMAN, ADRIAAN
Applicants: NCR CO, NCR CORPORATION
Application year: 1984
Main classification: H04L27
IPC codes: H04L27