OSCILLATOR CIRCUIT

Abstract: According to a oscillator circuit of the present invention, there is provided a negative switched capacitor circuit (10) having negative equivalent resistance, the value of which is determined according to the reference frequency and a frequency to be compared, and a positive switched capacitor circuit (20) having positive equivalent resistance, the value of which is determined according to the reference frequency. A constant DC voltage (V) is supplied in parallel to one terminal of the two switched capacitor circuits. The respective terminals of the switched capacitor circuits are commonly connected in order to produce the composite current of both output currents of the two switched capacitor circuits. The composite current is integrated by an integrator (30). Further, there is provided a Schmitt-type oscillating circuit (83). The oscillating frequency signal from the Schmitt-type oscillating circuit is supplied to the negative switched capacitor circuit. The higher level threshold voltage (V H ) of the Schmitt-type oscillating circuit (83) is determined according to the composite current of both output currents of the two switched capacitor circuits, thereby adjusting the frequency of the oscillating signal of the Schmitt-type oscillating circuit (83). The oscillating signal is supplied to the positive switched capacitor circuit. The lower level threshold voltage (V L ) of the Schmitt-type oscillating circuit is determined according to the composite current of both output currents of the two switched capacitor circuits. Therefore, the frequency of the oscillating signal of the Schmitt-type oscillating circuit is adjusted.

Description: The present invention relates to an oscillator circuit of the frequency locked loop type. Recently, devices which perform speech synthesis using digital technology have been developed and used. In such a device, an impulse and a white noise are used as a sound source. The signal from the sound source is passed through several digital filters to obtain an audio signal. The conditions for the digital filters depend on the audio signal to be achieved. Furthermore, the conditions for the digital filters in the digital speech synthesizer are set by analyzing and recognizing an actual voice. Fig. 1 is a conventional block diagram of a speech recognition circuit. In the figure, a mike amplifier 1 amplifies an analog signal from a microphone (not shown). The output from the mike amplifier 1 is supplied in parallel to, for example, four band-pass filters (BPF) 2A, 2B, 2C and 2D. The signals which pass through the four band-pass filters 2A, 2B, 2C and 2D are detected by four detectors, respectively. The detected four signals are supplied to four low-pass filter circuits 4A, 4B, 4C and 4D, respectively. The signals which pass through the low-pass filter circuits 4A, 4B, 4C and 4D are selectively supplied to an analog/digital converter (A/D) 6 through a multiplexer 5. The digital output from the analog/digital converter 6 results in a recognition result for the voice input through the microphone. The employment of switched-capacitor filter technology introduces higher integration and higher accuracy to the recent speech recognition circuit. In other words, the mike amplifier 1, band-pass filter circuits 2A through 2D and low-pass filter circuits 4A through 4D all adopt the switched capacitor circuit. For the control of the switched capacitor circuit, any circuit which uses a switched capacitor circuit is required to include an oscillator and a clock generator which produces various clock pulses from the oscillator output. When an accurate oscillating frequency is required, the actual oscillating frequency can be matched to the accurate oscillating frequency by knowing whether the actual oscillating frequency is greater or smaller than the accurate oscillating frequency, or the fluctuation or ratio between the actual oscillating frequency and the accurate oscillating frequency. A PLL (phase locked loop) circuit has been known as a means for accurately detecting which frequency is greater or smaller, or the fluctuation or ratio between the actual and accurate frequencies. However, the PLL circuit is not adapted to integration due to its complicated construction. Therefore, with prior art circuits with simplified constructions, an accurate oscillating frequency can not be obtained, thereby degrading the switched capacitor circuit which leads to a speech recognition circuit with poor accuracy. Prior art document "IEEE Journal of Solid-State Circuits", Volume SC-17, No. 4, August 1982, pages 775 to 778, New York, US, discloses a switched-capacitor frequency control loop comprising a resistor in parallel with a switched capacitor network, DC supply means for supplying a current to the resistor and the switched capacitor network, an integrator for integrating the composite current from the resistor and the switched capacitor network and a voltage controlled oscillator receiving the integrator output signal through a filter and feeding a two phase clock generator which controls the switches of the switched capacitor network. The switched capacitor network removes a charge from the integrator input, contrary to the resistor through which a charge is supplied to the integrator input. The resistor can be replaced by a switched capacitor network clocked by a constant frequency and supplying a current to the integrator input, thus having a positive equivalent resistance, dependent on the constant frequency. Further, prior art document GB-A-1 409 104 discloses an oscillator circuit of the voltage or current controlled oscillator type. Such voltage or current controlled oscillator is suitable to be used in a frequency controlled loop. Prior art document FR-A-2 441 174 describes a phase and frequency comparator where in a frequency control loop a frequency divider is used in the feedback loop of the oscillator, the output of the frequency divider feeding one input of the frequency comparator. Finally, prior art document Patents Abstract of Japan, Volume 7, No. 23 (P-171) (1168), January 29, 1983 (= JP-A-57175964) discloses a frequency detector in which a resistor and a capacitor are connected in parallel between the input and output terminals of an operational amplifier. It is an object of the invention to provide an oscillator circuit using a frequency comparing circuit which can stably generate with high accuracy an arbitrary multiple of frequencies from the reference signal derived from an oscillating signal, which is supplied from an oscillator such as a crystal oscillator which stably oscillates. To solve this object the present invention provides an oscillator circuit as specified in claim 1. Other objects and features of the present invention will be apparent from the following description taken in connection with the accompanying drawings in which: Fig. 1 is a block diagram showing a conventional construction of a speech recognition circuit; Fig. 2 is a circuit diagram showing a frequency comparing circuit, Fig. 3 is a circuit diagram showing another frequency comparing circuit; Fig. 4 is a detailed circuit diagram of a switched capacitor circuit 10 used in the frequency comparing circuit of Figs. 2 and 3; Fig. 5 is a detailed circuit diagram of a switched capacitor circuit 20 used in the frequency comparing circuit of Figs. 2 and 3; Figs. 6A and 6B are timing charts of the switched capacitor circuit shown in Fig. 4 or 5; Fig. 7 is a circuit diagram for application of the comparing circuit; Fig. 8 is a circuit diagram showing another frequency comparing circuit; Fig. 9 is a characteristic of the frequency comparing circuit shown in Fig. 8; Fig. 10 is a circuit diagram for application of the frequency comparing circuit; Figs. 11A and 11B are circuit diagrams showing an embodiment of an oscillator circuit; Fig. 12 is a waveform explaining an operation of the Schmitt trigger shown in Figs. 11A and 11B; Fig. 13 is a characteristic of an oscillating circuit shown in Figs. 11A and 11B; Fig. 14 is a block diagram showing another embodiment of an oscillator circuit; Fig. 15 is a detailed circuit diagram of an embodiment of a frequency divider shown in Fig. 14; Fig. 16 is a block diagram showing an embodiment of the oscillator circuit of the invention; Fig. 17 is a detailed circuit diagram of an embodiment of a synchronizing circuit shown in Fig. 16; and Figs. 18A through 18H are timing charts of the circuitry in Fig. 17 in which: Fig. 18A shows the output signal of the Schmitt trigger; Fig. 18B, the output signal of the frequency divider 110; Fig. 18C, the output signal of the F/F 121 in Fig. 17; Fig. 18D, the output signal of F/F 122 in Fig. 17; Fig. 18E, the inverted output signal of F/F 121 in Fig. 17; Fig. 18F, the output signal of NOR gate 124 in Fig. 17; Fig. 18G, the inverted output signal of NAND gate 127 in Fig. 17; and Fig. 18H, the output of NOR gate 128 in Fig. 17. Fig. 2 is a circuit diagram showing a frequency comparing circuit. The switched capacitor circuit 10 comprises a capacitor 11 and four switches 12 through 15. The circuit 10 has a negative equivalent resistance, and its value is determined corresponding to the frequency f s of an oscillating signal S s from an oscillator. One terminal of each switch 12 and 13 is connected to one terminal of the capacitor 11, and the other terminal of the switch 12 is grounded. The other terminal of the capacitor 11 is connected to one terminals of the switches 14 and 15, and the other terminal of the switch 14 is grounded. Among the four switches 12 through 15, a pair of switches 12 and 15 and a pair of switches 13 and 14 are alternatively turned on in response to the Signal S s . The switched capacitor circuit 20 comprises a capacitor 21 and four switches 22 through 25 and has a positive equivalent resistance, the value of which is determined corresponding to a reference signal S c with frequency f c . One terminal of the capacitor 21 is connected to one terminal of each switch 22 and 23, and the other terminal of the switch 23 is grounded. The other terminal of the capacitor 21 is connected to one terminal of each switch 24 and 25, and the other terminal of the switch 25 is grounded. A switch pair 22 and 24 and another switch pair 23 and 25 are alternatively turned on in response to a signal S c . The other terminal of the switch 13 in the switched capacitor circuit 10 and the other terminal of the switch 22 in the switched capacitor circuit 20 are connected at a common connection a. Futhermore, the positive electrode of a DC power source V is connected to the common connection a. The negative electrode of the DC power source V is grounded. Also, the other terminals of the switch 15 in the switched capacitor circuit 10 and the switch 24 in the switched capacitor circuit 20 are connected to a common connection b. Furthermore, an input terminal of an integrator 30 is connected to the common connection b. The integrator 30 is provided with a differential amplifier 31 having an inverted input terminal, a non-inverted input terminal and output terminal, and a capacitor 32. The capacitor 32 is connected between the inverted input terminal and output terminal, and the non-inverted input terminal of the differential amplifier 31 is grounded. The differential amplifier 31 operates at a voltage between a positive electrode voltage V DD and a negative electrode voltage V SS . The ground voltage is set at an intermediate potential between the positive and negative voltage, for example, at 0 V. The inverted input terminal of the differential amlifier 31 serves as the input terminal of the integrator 30. The integrator 30 produces on its output terminal a signal OUT corresponding to the magnitude relation between two frequencies f s and f c . The equivalent resistance R₁ of the switched capacitor circuit 10 (hereafter referred to as SC circuit) is given by: wherein C₁ represents the capacitance of the capacitor 11. As well, the equivalent resistance R₂ of the SC circuit 20 is given by: wherein C₂ represents the capacitance of the capacitor 21. As the one terminal of each two SC circuits 10 and 20, namely, the other terminals of the switches 13 and 22 are connected through the connection a to the positive electrode of the DC power source V, the positive bias from the DC power source V is supplied to both SC circuits 10 and 20. Therefore, a predetermined DC current flows into the respective SC circuits 10 and 20. As the SC circuit 10 has negative equivalent resistance R₁, the direction of the current I₁ is to the left in Fig. 2. Also, the SC circuit 20 has a positive equivalent resistance R₂, and the direction of the current I₂ is to the right in Fig. 2. In other words, the directions of the currents I₁ and I₂ are opposite to each other. A composite current of I₁ and I₂ is supplied to the integrator 30. When the composite current is 0, the output signal OUT of the integrator 30 becomes ground voltage. When I₁ > I₂ and current corresponding to the difference between I₁ and I₂ flows from the integrator 30, the level of the output signal OUT of the integrator 30 becomes high (V DD level). On the contrary, when I₂ > I₁ and current corresponding to the difference between I₁ and I₂ flows into the integrator 30, the level of the output signal OUT of the integrator 30 become low (V SS level). Suppose that the capacitance C₁ of the capacitor 11 in the SC circuit 10 and the capacitance C₂ of the capacitor 21 in the SC circuit 20 are equally set. When the frequency f s of the oscillating signal S s coincides with the frequency f c of the reference signal S c , the output signal OUT of the integrator 30 becomes ground voltage. Furthermore, when the frequency f s is greater than the frequency f c , the level of the output signal OUT of the integrator 30 becomes high. On the other hand, when the frequency f c is smaller than the frequency f c , the level of the output signal OUT of the integrator 30 becomes low. In summary, the output signal OUT of the integrator 30 varies its level according to the cyclical relationship of the two frequencies f s and f c . Therefore, the comparison of the two frequencies can be performed by determining the output signal OUT. The equivalent resistances R₁ and R₂ of the SC circuits 10 and 20 are determined only by the frequencies f s and f c , provided that the capacitances C₁ and C2 are constant as shown in the equations (1) and (2). Since the capacitances C₁ and C₂ can be set with a relatively higher accuracy than the resistances, the respective frequencies f s and f c can be converted into the resistances R₁ and R₂ with high accuracy. Furthermore, the integrator 30 achieves the signal OUT by integrating the composite values of flowing currents corresponding to the resistances R₁ and R₂. Thus, the comparison between f s and f c can be performed with high accuracy even if the slew rate or gain of the differential amplifier 31 fluctuates. As described above, the comparison and detection can be performed with high accuracy on the cyclical relationship between the frequency f s of the oscillating signal S s from the oscillator and the frequency f c of the reference signal S c . Therefore, the frequency f s can be matched with the reference frequency f c with high accuracy by using the output signal OUT. Also in this comparing circuit, the apparent frequency of the reference signal S c can be varied from the actual value f c by changing the capacitances C₁ and C₂ of the capacitors 11 and 21 in the SC circuits 10 and 20. For example, by setting the values of the capacitors 11 and 21 to satisfy C₁ = 2C₂, the comparison and detection of the cyclical relationship between 2f c and f s can be performed. Fig. 3 is a circuit diagram showing a construction of another frequency comparing circuit. The circuit in fig. 3 is same as that in Fig. 2 except that a resistor 33 is connected in parallel to the capacitor 32 in the integrator 30. In this comparing circuit, as in the comparing circuit of Fig. 3, when the composite value of the output currents I₁ and I₂ in the SC circuits 10 and 20 is 0, the output signal OUT is set to ground voltage. However, when the composite value is not 0, the output signal OUT is set to less than the V DD or V SS level in accordance with the electrode of the composite current. Figs. 4 and 5 show detailed circuits of the SC circuits 10 and 20 used in the above comparing circuit respectively. The same numerals as in Figs. 2 and 3 are used for the same parts in Figs. 4 and 5. Furthermore, two signals S s1 and S s2 or S c1 and S c2 are used whose phases are different than each other as shown in the timing charts of Figs. 6A and 6B. The switches 12 through 15 in the SC circuit 10 having negative resistance comprise, as shown in Fig. 4 CMOS switches 52 through 55 in which N channel MOSFETs 41 through 44 and P channel MOSFETs 45 through 48 are connected in parallel to each other. The signal S s1 in Fig. 6A is supplied to the gates of the N channel MOSFETs 41 and 44 and through a CMOS inverter 49 to the gates of the P channel MOSFETs 45 and 48. The signal S s2 in Fig. 6B is supplied to the gates of the N channel MOSFETs 42 and 43 and through a CMOS inverter 50 to the gates of the P channel MOSFETs 46 and 47. Now the switching control of the CMOS switches 52 through 55 will be described in response to the signals S s1 and S s2 under the condition that the DC voltage V1 is supplied to the other terminal of the CMOS switch 53 and the other terminal of the CMOS switch 55 is grounded. When the level of the signal S s2 is high, the CMOS switches 53 and 54 turn on. Then, the other terminal (point C in Fig. 4) of the capacitor 11 will be charged at -C·V₁. When the level of the signal S s1 is high, the CMOS switches 52 and 55 turn on. At this time, the positive charge +C₁·V₁ from the ground is supplied through the CMOS switch 55 to the point C so as to cancel the negative charge at the point C. When such an operation is repeated f c times per second and the direction of the current flowing through the CMOS switch 55 from the point C is positive, the value of the current I which flows in the SC circuit is given by: -I = C₁·V₁·f s (3) The equivalent resistance R in the SC circuit 10 is obtained by dividing the current I by the supply voltage V1. Thus, the equivalent resistance R is given by: The right side in the equation (4) is the same as the right side in equation (1). Therefore, the equation (4) reveals that the SC circuit 10 in Fig. 4 has the negative equivalent resistance corresponding to the frequency f s . The switches 22 through 25 in the SC circuit 20 comprise the CMOS switches 72 through 75 in which N channel MOSFETs 61 through 64 and P channel MOSFETs 65 through 68 are connected in parallel, respectively, as shown in Fig. 5. The signal S c1 in fig. 6A is supplied to the gates of the N channel MOSFETs 61 and 63 and through the CMOS inverter 69 to the gates of the P channel MOSFETs 65 and 67. The signal S c2 in Fig. 6B is supplied to the gates of the N channel MOSFETs 62 and 64 and through a CMOS inverter 70 to the P channel MOSFETs 66 and 68. Now a switching control will be described of the CMOS switches 72 through 75 in response to the signals S c1 and S c2 under the condition that the DC voltage V₂ is supplied to the other terminal of the CMOS switch 72 and the ground voltage is supplied to the other terminal of the CMOS switch 74. When the level of the signal S c1 is high, the CMOS switches 72 and 74 turn on. At this time, the capacitor 21 is charged to C₂·V₂. Then, when the level of the signal S c2 becomes high, the CMOS switches 73 and 75 turn on. Therefore, the charge stored in the capacitor 21 is discharged. When such an operation is repeated f c times per second, and the direction of the current flowing from the other terminal of the capacitor 21 (point d in Fig. 5) through the CMOS switch 74 to the ground is positive, the current I which flows in the SC circuit 20 is given by: I = C₂·V₂·f c (5) The equivalent resistance R in the SC circuit is obtained by dividing the supply voltage V₂ by the current I. That is: The right side in equation (6) is the same as that in equation (2). Thus, the equation (6) reveals that the SC circuit 20 in Fig. 5 has a positive equivalent resistance corresponding to the frequency f c . Fig. 7 is a circuit diagram showing an applied construction of the comparing circuit. The circuit of this application comprises two frequency comparing circuits 100 and 200 constructed like those in Fig. 2 and an AND gate 300 which receives both the output signals OUT1 and OUT2. The SC circuit having the negative equivalent resistance (referred to by 10 in fig. 2) in one frequency comparing circuit 100 is controlled by a signal S ca having a higher reference frequency f ca . The SC circuit having a positive equivalent resistance (referred to by 20 in Fig. 2) is controlled by a signal S s having a frequency f s to be compared. The SC circuit having a negative equivalent resistance (referred by 10 in Fig. 2) in the frequency comparing circuit 200 is controlled by a signal S cb having a lower frequency f c and the SC circuit having a positive equivalent resistance is controlled by the signal S s . In this circuit, it is assumed that all the capacitances in the SC circuit are equally set. The level of the output signals OUT1 and OUT2 of the frequency comparing circuits 100 and 200 becomes high, thus the output level of the AND gate 300 becomes high only when the relation f ca > f s > f cb is satisfied between the frequencies f ca , f cb and f s of the signals S ca , S cb and S s , respectively. In other words, the circuit can detect whether the frequency f s to be compared is between the higher reference frequency f ca and the lower reference frequency signal f cb . In the comparing circuits in Figs. 2 and 3, a common DC power source V is used to flow DC current in the SC circuits 10 and 20. However, different DC power sources may be used for the SC circuits 10 and 20. When using different DC power sources, the apparent frequency of the reference signals S c can be varied from the actual frequency f c by using different DC power sources as changing values of the capacitors 11 and 21. In the above comparing circuits, one SC circuit 10 having a negative equivalent resistance is controlled by the signal S s , and the other SC circuit 20 having a positive equivalent resistance is controlled by the signal S c . However, the signals S s and S c are interchangeable. Fig. 8 shows a frequency comparing circuit which can perform fluctuations between the two signal's frequencies with high accuracy. The same numerals are used for the same parts in Fig. 2 and the descriptions are omitted. In this frequency comparing circuit, two resistors 76 and 77 are connected in series between the positive electrode power source V DD and the ground. The other terminal of the switch 13 in the switched capacitor circuit 10 is connected to a serial connection a1 between the resistors 76 and 77. Furthermore, two resistors 78 and 79 are serially connected between V DD and the output point OUT of voltage V₀. The other terminal of the switch 22 of the SC circuit 20 is connected at the serial connection a2 between the resistors 78 and 79. The differential amplifier 31 operates at a voltage between the positive electrode power source voltage V DD and negative electrode power source voltage V SS whose absolute value is equal that of V DD . The ground potential is set between V DD and V SS , for example, to 0 V. In the integrator 30, the inverted input terminal of the differential amplifier 31 serves as the input terminal, and from the output terminal, voltage proportional to the fluctuation between the two frequencies f s and f c is provided. In such a construction, a constant positive voltage V₁, determined by the resistance ratio of the resistors 76 and 77, and the voltage V DD is supplied to the connection a1 where the one terminal of the SC circuit 10, that is, the other terminal of the switch 13, is connected. The equivalent resistance R₁ of the SC circuit 10 becomes negative as in the equation (1) if the value of the capacitor 11 is C₁. Thus, the direction of the current I₁ flowing into the SC circuit 10 is to the left in Fig. 8. Positive or negative voltage V₂, determined by the resistance ratio of the resistors 78 and 79, voltage V DD , and the output voltage V₀ from the integrator 30, is supplied to the connection a2 where one terminal of the SC circuit 20, that is, the other terminal of the switch 22, is connected. The equivalent resistance R₂ of the SC circuit 20 becomes positive as in the equation (2) if the value of the capacitor 21 is C₂. When the voltage V₂ is positive, the direction of the current I2 flowing into the SC circuit 20 is to the right in Fig. 8. The integrator 30 integrates the composite current of I₁ and I₂ and reduces voltage V₀. Now supposed that the capacitance C₁ of the capacitor 11 in the SC circuit 10 is equal to the capacitance C₂ of the capacitor 21 in the SC circuit 20, all the resistors 76, 77, 78 and 79 have the same resistance, and the voltages V1 and V2 are equal. Under this condition, when the frequency f s of the oscillating signal S s coincides with the frequency f c of the reference signal S c1 , current I₁ is equal to current I2, and their directions are opposite to each other as shown in Fig. 8. Therefore, the composite current of I₁ and I₂ become 0, and the output voltage of the integrator 30 becomes ground voltage. When the output voltage V₀ is ground voltage, the voltage V₂ at the connection a2 becomes (V DD -0)/2 as the values of the resistors 78 and 79 are equal. Also, the voltage V₁ at the connection a1 becomes (V DD -0)/2 as the values of the resistors 76 and 77 are equal. Therefore, voltage V₁ is equal to voltage V₂. Thus, the output voltage V₀ of the integrator 30 becomes ground voltage (0 V) when the frequency f s coincides with the frequency f c . When the output voltage V₀ is equal to the ground voltage, the voltage V₂ at the connection a2 becomes (V DD -0)/2 as the values of the resistors 78 and 79 are equal. Also, the voltage V₁ at the connection a1 becomes (V DD -0)/2 as the values of the resistors 76 and 77 are equal. Therefore, voltage V₁ is equal to voltage V₂. Thus, the output voltage V₀ of the integrator 30 is fixed at the ground level when the frequency f s coincides with the frequency f c . On the other hand, when the frequency f s is lower than the frequency f c , the equivalent resistance of the SC circuit 10 given by the equation (1) becomes greater. Since the value of voltage V₁ supplied to the SC circuit 10 is constant, the greater the resistance R₁, the smaller the current value I₁. As a result, the composite current of I₁ and I₂ becomes positive (in the direction of flowing into the integrator 30). The integrator 30 integrates the composite current and produces negative voltage V₀ proportional to the current. The voltage V₀ serves to lower the voltage V₂ at the connection a2, so that the value of the current flowing in the SC circuit 20 becomes smaller. When the current I₁ and I₂ are balanced, the output voltage V₀ of the integrator 30 becomes constant. Now when the frequency f s rises from the balance and becomes higher than the frequency f c , the equivalent resistance R₁ of the SC circuit 10 becomes smaller. Thus, the value of the current I₁ flowing in the SC circuit 10 becomes greater. As a result, the composite current of I₁ and I₂ become negative (in the direction of flowing out of the integrator 30). At this time, the integrator 30 integrates the composite current and produces a positive voltage V₀ proportional to the current value. Thus produced voltage V₀ serves to raise the voltage V₂ at the connection a2. Therefore, the value of current I₂ flowing in the SC circuit 20 becomes greater. When the balance between the currents I₁ and I₂ is obtained, the output V₀ of the intergrator 30 becomes constant. Fig. 9 shows a characteristic of the frequency comparing circuit in Fig. 8 when C₁ = C₂. In the figure, the frequency f s of the oscillating signal S s is measured on the abscissa, and the output voltage V₀ of the integrator 30 is measured on the ordinate. As is shown, there is a proportional relation between the frequency f s and the output voltage V₀. Therefore, from the output voltage V₀, the fluctuation between two frequencies f s and f c can be measured. That is, when the output voltage V₀ is 0, there is no fluctuation between the frequencies. When the voltage V₀ is positive, the frequency f s fluctuates corresponding to a voltage higher than the frequency f c , and when the voltage V₀ is negative, the frequency f s fluctuates corresonding to a voltage lower than the frequency f c . Furthermore, the equivalent resistances R₁ and R₂ in the SC circuits 10 and 20 depend, as is shown in the equations (1) and (2), only on the frequencies f s and f c if the values of C₁ and C₂ are constant, respectively. Therefore, the values of the capacitors C₁ and C₂ can be set with relatively higher accuracy compared to those of the resistors. Thus, the frequencies f s and f c can be converted into equivalent resistances R₁ and R₂ with higher accuracy. Also, in this frequency comparing circuit, the apparent frequency of the reference signal S c can be varied from the actual frequency f c by setting values C₁ and C₂ of the capacitors 11 and 21 and values of resistors 76 and 79. For example, by setting the values of the capacitors 11 and 21 to satisfy C₁ = 2C₂, the gradient of the characteristic line in Fig. 9 will be doubled. In this case, V₀ becomes 0 at the point of fs = f c /2 . Therefore, the output voltage V₀ corresponding to the fluctuation between the half-divided frequency of the frequency f c and frequency f s can be obtained. If the value of resistor 76 is set to be equal to that of resistor 77 and the value of resistor 78 is set to be greater than that of resistor 79, there is a relation of V₁ > V₂ at V₀ = 0. If the value of resistor 78 is set three times that of resistor 79, the relations are V₁ = V DD /2 and V2 = (V DD -V₀)/4 . Therefore, the gradient of the characteristic line in Fig. 9 will be doubled so that the same effect can be obtained as C₁ = 2C₂. Using the circuit, the applicant has conducted an experiment where 100PF is selected for C₁ and C₂, +7V for V DD , -7V for V SS and 2KHz, 4KHz and 8KHz for f c , respectively. As a result of the experiment, the output voltage V₀ represents linear characteristic when the frequency f s is in the range of 0.1fc through 1.7 or 1.8fc. Fig. 10 is a circuit diagram showing an applied construction of the frequency comparing circuit. The application circuit is constructed in such a way that the output voltage V₀ of a frequency comparing circuit 400, constructed as in Fig. 8, is supplied to an inverter 500, thereby to obtain a logic signal OUT with either a high or low level set according to the voltage V₀. Figs. 11A and 11B are circuit diagrams showing a construction of an embodiment of the oscillator circuit using the frequency comparing circuit device in Fig. 10. In the figures, the same numerals are used for the same parts in Fig. 2, thereby to omit the description thereof. In the figure, the output of the integrator 30 is supplied through a voltage buffer circuit 80 to a Schmitt-type oscillator 83. The voltage buffer circuit 80 comprises a differential amplifier 81 with an inverted input terminal, non-inverted input terminal and output terminal. The inverted input terminal is connected to the output terminal. The inverted input terminal is supplied with the voltage V₀. The Schmitt-type oscillator 83 is provided with a differential amplifier 84 with a non-inverted input terminal, inverted input terminal and output terminal. A higher level threshold voltage V H and lower-level threshold voltage V L are alternatively supplied to the non-inverted input terminal of the differential amplifier 84. The higher level threshold voltage V H is obtained at the serial connection c of the two resistors 85 and 86 serially connected at the point between the positive power supply voltage V DD and the output terminal of the voltage buffer circuit 80. The voltage V H is input to the non-inverted input terminal of the differential amplifier 84 through a CMOS switch 89 which comprises the P-channel MOSFET 87 and N-channel MOSFET 88. The lower level threshold voltage V L is obtained at the serial connection d of the resistors 90 and 91 serially connected between the point at which the positive power supply voltage V DD is applied and the point at which the negative power supply voltage V SS is applied. The voltage V L is input to the non-inverted input terminal of the differential amplifier 84 through a CMOS switch 94 which comprises the P channel MOSFET 92 and N channel MOSFET 93. Furthermore, DC voltage V V1 is applied to the point a1 of the switched capacitor circuit 10, and variable DC voltage V V2 is applied to the point a2 of the switched capacitor circuit 20. Two inverters 95 and 96 are cascade-connected to the output terminal of the differential amplifier 84, and a resistor 97 and a capacitor 98 are serially connected between the output terminal of the inverter 96 in the lower stream and the point at which the voltage V SS is applied. The output signal of the inverter 96 is input in parallel to a gate of an N-channel MOSFET 88 which constitutes the CMOS switch 89 and a gate of a P-channel MOSFET 92 which constitutes the CMOS switch 94 as well as to two inverters 99 and 101. The output signal of the inverter 99 is supplied in parallel to a gate of a P-channel MOSFET 87 which constitutes the CMOS switch 89 and a gate of an N-channel MOSFET 93 which constitutes the CMOS switch 94. Furthermore, a voltage at the point e where the resistor 97 and the capacitor 98 are serially connected is input to the inverted input terminal of the differential amplifier 84. The output signal of the inverter 101 is supplied to another circuit as the oscillating output signal S s and is fed back to the SC circuit 10 through a line 102. In this embodiment, the three differential amplifiers 31, 81 and 84 operate at a voltage between the positive power supply voltage V DD and the negative power supply voltage whose absolute value is equal to V DD . The voltage at the ground is set to 0 V which is intermediate between V DD and V SS . Now the operation of the Schmitt-type oscillating circuit 83 in the embodiment will first be described. Suppose that the resistance ratio of the resistors 85 and 86 and the threshold voltage V H obtained at the point c corresponding to the output voltage V₀ from the voltage buffer amplifier 80 are fixed, and that the voltage V H is higher than the threshold voltage V L obtained at the point d corresponding to the resistance ratio of the resistors 90 and 91. If the level of the output signal of the differential amplifer 84 is high (V DD level), the level of the output signal of the inverter 96 becomes high, so that the capacitor 98 discharges through the resistor 97 at the time constant τ corresponding to the resistor 97 and the capacitor 98. Thus, the voltage at the point e raises gradually. On the other hand, when the level of the output signal of the inverter 96 is high, the CMOS switch 89 turns on. Therefore, a high level threshold voltage V H is input to the non-inverted input terminal of the differential amplifier 84. The differential amlifier 84 compares the threshold voltage V H and the voltage at the point e. When the voltage at the point e reaches the threshold voltage V H , the output signal of the differential amplifier 84 is inverted from a high level to a low level (V SS level). After the level inversion, the output signal of the inverter 96 is also inverted into two levels, and the capacitor 98 which has been charged now discharges with the time constant τ to the V SS level through the resistance 97. Therefore, the voltage at the point e then lowers gradually. On the other hand, when the output signal of the inverter 96 is at a low level, the CMOS switch 94 turns on and the lower level threshold voltage V L is input to the non-inverted input terminal of the differential amplifier 84 through the CMOS switch 94. Thus, the differential amplifier 84 compares the threshold value V L with the voltage at the point e. As a result, the lowering voltage at the point e reaches the threshold value V L , and the output signal of the differential amplifier 84 is again inverted to a high level. As described above, the differential amplifier 84 alternatively compares the voltage at the point e with the threshold voltage V H and the threshold voltage V L . Therefore, the output signal of the inverter 96 becomes an oscillating signal in which a high level and a low level are repeated alternatively. Futhermore, the output signal S s of the inverter 101 is also the oscillating signal as it is the inverted signal through the inverter 96. The frequency f s of the oscillating signal S s of the Schmitt-type oscillator 83 is constant when the threshold voltages V H and V L are constant. The value f s corresponds to the Schmitt width (|V H | + |V L |) determined by the threshold voltages V H and V L , and the time constsnt τ is determined by the resistor 97 and the capacitor 98. Fig. 12 is a waveform showing the relation of the voltage Ve at the point e and the output voltage S s of the inverter 96. On the other hand, the equivalent resistance R₁ of the SC circuit 10, to which the oscillating signal S s from the Schmitt-type oscillating circuit 83 is input, is given by the equation (1) when the value of the capacitor 11 is C₁, as shown in Fig. 2. Similarly, the equivalent resistance R₂ of the SC circuit 20, to which the signal S c is input, is given by the equation (2) when the value of the capacitor 21 is C₂. As the positive voltage from the DC power supply V₁ or variable DC power supply V₂ is supplied to one terminal of the SC circuits 10 and 20, that is, the other terminal of the switch 13 in the SC circuit 10 and the other terminal of the switch 22 in the SC circuit 20, direct current flows in the SC circuits 10 and 20, respectively. On the other hand, as the SC circuit 10 has a negative equivalent resistance R₁, the direction of the current I₁ flowing in the SC circuit 10 is to the left (negative direction) in Figs. 11A and 11B. And as other SC circuit 20 has a positive equivalent resistance R₂, the direction of the current I₂ flowing in the SC circuit 20 is to the right (positive direction) in Figs. 11A and 11B. In other words, the directions of I₁ and I₂ are opposite to each other. The composite current of I₁ and I₂ is supplied to the integrator 30. When the composite current is negative, that is, the current I₁ is greater than the current I₂, the output voltage V₀ of the integrator 30 rises gradually to the positive power supply voltage V DD . On the contrary, when the composite current of I₁ and I₂ is positive, that is, the current I₂ is greater than the current I₁, the output voltage V₀ of the integrator 30 lowers gradually toward the negative power supply voltage. When the output currents I₁ and I₂ of the SC circuits 10 and 20 are balanced and their composite current becomes 0 during rising and lowering of the output voltage V₀ of the integrator 30, the change of the output voltage V₀ is ceased and can never be changed thereafter. Supposed that the value of the DC power supply V₁ and the value of the variable DC power supply V₂ are equally set, and the values C₁ and C₂ of the capacitors 11 and 12 in the SC circuits 10 and 20 are equally set. Then, in the SC circuits 10 and 20 the composite current of I₁ and I₂ becomes 0, and the output voltage V₀ of the integrator 30 is fixed to some value when the coincidence between the frequecies f s and f c of the signals S s and S c is achieved. When value V₀ is fixed, the higher threshold voltage V H formed in the Schmitt-type oscillator 83 is fixed, thereby fixing the Schmitt width (|V H | + |V L |) and frequency f s of the oscillating signal S s . Now suppose the frequency f s lowers from the balance between the two frequencies f s and f c . When the frequency f s lowers, the equivalent reistance R₁ of the SC circuit 10 given by the equation (1) becomes greater than before, and thus the absolute value of the current I₁ becomes smaller. The composite current of I₁ and I₂ becomes positive, and the output voltage V₀ of the integrator 30 lowers towards V ss . Then, the voltage V H obtained at the point c in the Schmitt-type oscillator lowers. Therefore, the Schmitt width (|V H | + |V L |) is narrower, thereby raising the frequency f s of the oscillating signal S s in the Schmitt-type oscillating circuit 83. When frequency f s rises, the equivalent resistance R₁ of the SC circuit 10 becomes smaller and the absolute value of the current I₁ gradually becomes greater. As a result, the composite current of I₁ and I₂ decreases and becomes 0 when the frequency f s coinciding with the frequency f c lowers, and the output voltage V₀ of the integrator 30 rises toward V DD until f s equals f c . When the output voltage V₀ returns to its original value, the rise of the voltage is ceased, and the rise of frequency f s is also ceased when the frequency f s of the oscillating signal S s coincides with the frequency f c . Now suppose the frequency f s rises under the condition that the frequency f s coincides with the frequency f c . When the frequency f s rises, the equivalent resistance R₁ of the SC circuit 10 given by the equation (1) becomes lower, and the absolute value of the current I₁ becomes greater. The composite current of I₁ becomes greater. The composite current of I₁ and I₂ becomes negative, and the output voltage V₀ of the integrator 30 rises toward V DD . When the voltage V₀ rises, the higher threshold voltage V H becomes higher than before, and the Schmitt width (|V H | + |V L |) becomes wider than before. Therefore, the frequency f s of the oscillating output signal S s in the Schmitt-type oscillating circuit 83 lowers. When the frequency f s lowers, the equivalent resistance R₁ of the SC circuit 10 becomes greater, and the absolute value of the current I₁ gradually becomes smaller. The composite value of the current I₁ and I₂ increases toward 0 from negative and becomes 0 when f s coincides with f c . Thus, when the frequency rises, the output voltage V₀ of the integrator 30 lowers toward V ss until f s coincides with f c . When the output voltage V₀ becomes its original value, the lowering of the voltage ceases. And lowering of the frequency f s is also ceased when the frequency f s of the oscillating output signal S s coincides with f c . In other words, in this embodiment, the frequency f s of the oscillating output signal S s is controlled so as to coincide with the frequency f c of the signal S c . Now the setting of the variable DC power supply V2 will be explained. When the voltage V₂ is set higher than V₁, the output current I₂ of the SC circuit 20 becomes greater than before since the equivalent resistance R₂ of the SC circuit 20 is constant. In order to balance the currents I₁ and I₂ and to make the composite current 0, the absolute value of the output current I₁ of the SC circuit 10 must be greater than before. When the current I₂ becomes greater, the output voltage V₀ of the integrator 30 lowers towards V ss , and thereafter, the frequency f s of the oscillating output signal of the Schmitt-type oscillator 83 rises. The rise in the frequency f s continues until the absolute value of the output current I₁ of the SC circuit 10 coincides with the absolute value of the current I₂. When the voltage V₂ is set higher than the voltage V₁, the frequency f s of the oscillating output signal S s is controlled to be constant at a point higher than the frequency f c . When the voltage V₂ is set lower than the voltage V₁, the output current I₂ of the SC circuit 20 becomes smaller than before. In order to balance the currents I₁ and I₂ and to make the composite current 0, the absolute value of the output current I₁ of the SC circuit 10 must be smaller than before. When the current I₂ becomes smaller, the output voltage V₀ of the integrator 30 rises toward V DD and, therefore the frequency f s of the oscillating output signals S s lowers. The lowering of the frequency f s continues until the absolute value of output current I₁ of the S c circuit 10 coincides with the absolute value of I₂. In other words, when V₂ is set lower than V₁, the frequency f s of the oscillating output signal S s is controlled to be lower than f c . In this embodiment, the condition where the oscillating frequency f s becomes stabilized is when the composite current I₁ and I₂ of the SC circuits 10 and 20 becomes 0. Thus, the stabilizing condition is give by: I₁ + I₂ = 0 (7) By substituting the relation of R₁ and R₂ given in the equations (1) and (2) for equation (7), the following equation is obtained: -C₁·f s ·V₁ + C₂·f c ·V₂ = 0 (8) From the equation (8), the following equation is obtained: As it is assumed that C₁ = C₂, f s equals f c times V₂/V₁. Therefore, the signal S s having the frequency f s equal an arbitrary multiple of the frequency f c . Fig. 13 shows a characteristic of the above embodiment when C₁ = C₂. In the figure, the frequency f s of the oscillating signal S s is measured on the abscissa and the value of the variable power supply V₂ is measured on the ordinate. As is shown, f s is related to V₂ linearly. The equivalent resistances R₁ and R₂ in the SC circuits 10 and 20 are determined only by the frequencies f s and f c if C₁ = C₂, as shown in the equations (1) and (2). Since the values C₁ and C₂ of the capacitors 11 and 21 can be set with relatively higher accuracy than the resistances, the frequencies f s and f c can be converted into resistances R₁ and R₂ with high accuracy. The difference between the resistances R₁ and R₂ is converted into current and then converted into voltage. The oscillating frequency f s of the Schmitt-type oscillator 83 can be adjusted corresponding to the above voltage. Thus, the frequency f s can be stably adjusted with high accuracy. In the above embodiment, the adjustment of the oscillating frequency f s is performed by varying the value of the variable DC power supply V₂. That is, as is apparent from the equation (8), the values C₁ and C₂ of the capacitors 11 and 21 in the SC circuits 10 and 20 are changed, and a multiple frequency corresponding to the ratio of C₁ and C₂ is produced. For example, by setting the C₁ and C₂ as n·C₁ = C₂ , the frequency f s can be adjusted to be a frequency n times as much as the frequency f c . Furthermore, the n must not be an integer but may be a real number including a decimal. This applies to when the adjustment of the frequency f s is performed by changing the variable DC power supply V₂. Furthermore, V₁ may be a variable power supply instead of V₂. Fig. 14 is a block diagram showing another embodiment of the oscillator circuit. In the embodiment, a frequency 110 is added to the embodiment of Figs. 11A and 11B. The frequency divider 110 divides the oscillating output signal S s of the Schmitt-type oscillator 83 into 1/n. The SC circuit 10 is controlled by the 1/n divided signal S' s in stead of the oscillating signal S s . According to the above construction, the frequency f s of the oscillating output signal S s is controlled to be constant at a frequency n times as much as the frequency f s in the embodiment of Figs. 11A and 11B. Fig. 15 is a circuit diagram showing an example of the frequency divider 110 in Fig. 14 in detail. The frequency divider 110 in this embodiment frequency divides the oscillating output signal by 1/8. The divider 110 is a known circuit and comprises three D-type flip-flops 111 through 113. Each data input terminal D of the flip-flops 111 through 113 is connected to the respective first output terminal Q , and each second output terminal Q in the previous stage is connected to each clock input terminal cp in the next stage. The oscillating output signal S s from the Schmitt-type oscillator 83 is supplied to the clock input terminal of the flip-flop 111 in the first stage and, the 1/8 frequency-divided signal S' s is output from the second output terminal of the flip-flop 113 in the final stage. The division ratio can be modified by adjusting the number of flip-flops. Fig. 16 is a block diagram showing an embodiment of the present invention. This embodiment further comprises the frequency divider 110 in Fig. 14 and two synchronizing circuits 120A and 120B. The frequency divider 110 is constructed the same as that in Fig. 15 and frequency divides the oscillating signal S s by 1/8 to produce the signal S' s . The synchronizing circuit 120A synchronizes the 1/8 frequency-divided signal S' s with the signal S s , which is not frequency divided. The SC circuit 10 is controlled by the output signal S' so of the synchronizing circuit 120A. The synchronizing circuit 120B synchronises the reference signal S c with the oscillating signal S s of the Schmitt-type oscillator 83. The SC circuit 20 is controlled by the output signal S co of the synchronizing circuit 120B. Fig. 17 shows the synchronizing circuits 120A and 120B in Fig. 16 in detail. The synchronizing circuits 120A and 120B are the same in construction but different in their input signals. The oscillating signal S s from the Schmitt-type oscillator 83 is supplied to the clock input terminals (cp) of the synchronized D-type flip-flops 121 and 122. The frequency-divided signal S′ s from the frequency divider 110 is supplied to the data input terminal D of the synchronizing circuit 120A, and the signal S c is supplied to the input terminal D of the synchronizing circuit 120B. The signal Q₁ of the output terminal Q of the one flip-flop 121 is supplied to the data input terminal of the other flip-flop 122 and to the inverter 123. The output signal Q ₁ of the inverter 123 is supplied in parallel together with the signal Q₂ from the other flip-flop 122 to a NOR gate 124. The output signal X of the NOR gate 124 is supplied to a NAND gate 127 and a NOR gate 128 through two cascade-connected inverters 125 and 126. The output signal of the inverter 126 is supplied to the NAND gate 127 and NOR gate 128 through four cascade-connected inverters 129 through 132. Furthermore, the output signal of the NAND gate 127 is supplied to the inverter 133. The output signal of the inverter 133 is supplied to the SC circuit 10 as the signal to control the switches 13 and 14 in the SC circuit 10 when the signal to be supplied to the flip-flop 121 is S′ s , and is supplied to the SC circuit 20 as the signal to control the switches 23 and 25 in the S c circuit 20 when the signal to be supplied to the flip-flop 121 is S c . The output signal of the NOR gate 128 is supplied to the SC circuit 10 as the signal to control the switches 12 and 15 in the SC circuit 10 when the signal to be supplied to the flip-flop 121 is S′ s and is supplied to the SC circuit as the signal to control the switches 22 and 24 in the SC circuit 20 when the control S c to be supplied to the flip-flop 121 is S c . The synchronizing circuits 120A and 120B as shown in Fig. 17 operate as follows. Suppose that the signal S′ s has the phase distortion as shown in Fig. 18B against the oscillating output signal S s (Fig. 18A) from the Schmitt-type oscillator 83. The flip-flops 121 and 122 set the levels of the output signal Q₁ and Q₂ in Figs. 18C and 18D to those of the input signals in synchronism with the leading edge of the clock input signal, that is, the signal S s , and holds the previous output level at the trailing edge of the signal Sc. Therefore, the output signals Q₁ and Q₂ vary as shown in Figs. 18C and 18D. Thus, the pulse signal is obtained as the output signal X (Fig. 18F) of the NOR gate 124, which is synchronized with the signal S s and is at a high level during one period of the signal S s . Using the delayed time obtained by the inverters 129 through 132. The circuit which comprises the inverters 129 through 132 to which the signal X is supplied (the inverter 133, NAND gate 127 and NOR gate 128) forms from the signal X a pair of signals S′ so1 and S′ so2 , as shown in Figs. 18G and 18H, corresponding to the two phase signals S s1 and S s2 , as shown in Figs. 6A and 6B, with high level durations which do not overlay each other. The switches 12 and 15 in the S c circuit 10 controlled by the signal S' so1 turn on for nearly one cycle of the signal S s . In other words, the duration the positive charge flows into the SC circuit 10 is set to the duration corresponding to nearly one cycle of the signal S s . Also in the synchronizing circuit 120B to which the signal S c is supplied as the input signal, a pulse signal is obtained which is in synchronism with the signal S s and is at a high level during one cycle of the signal S s as the output signal X of the NOR gate 124. Furthermore, a pair of signals S co1 and S co2 are formed corresponding to the two-phase signals S c1 and S c2 as shown in Figs. 6A and 6B. The switches 22 and 24 in the SC circuit 20 controlled by the signal corresponding to the signal S' so1 in Fig. 18G turn on for nearly one cycle of the signal S s . That is, the duration for the positive charge flowing out from the SC circuit 20 is set to nearly one cycle of the signal S s as in the case of the SC circuit 10. Therefore, the embodiment of Fig. 16 satisfies the equation (8), thereby preventing the generation of an oscillating frequency error when the oscillating frequency is relatively large. As shown in Figs. 18G and 18H, the high level durations of the signal S' so1 and S' so2 (or S' co1 and S' co2 ) are not overlayed. Therefore, the switches 12 and 15 and the switches 13 and 14 are not simultaneously turned on, and a difference between an ideal value and a measured value of the current I₁ is not generated. This applies also to the SC circuit 20. The present invention is not limited to the above embodiments but various modifications can be performed. For example, in the embodiments of Figs. 11A, 11B, 14 and 16, the oscillating circuit in which the oscillating frequency is controlled by the composite current of the output currents of the SC circuits 10 and 20 is a Schmitt-type oscillating circuit 83. However, any type of oscillating circuits having the same function may be used. Further in the above embodiment, the SC circuit 10 having negative equivalent resistance is controlled by the signal S s , and the S c circuit 20 having positive equivalent resistance is controlled by the signal S c . However, these signals are interchangeable.

Claims: An oscillator circuit, comprising: first means (10) comprised of a switched capacitor circuit and having negative equivalent resistance, the value of which is determined according to the frequency of a first signal: second means (20) comprised of a further switched capacitor circuit and having positive equivalent resistance, the value of which is determined according to the frequency of a second signal; third means (VV1,VV2) for supplying DC voltages to said first and second means (10, 20); fourth means (30) for integrating composite current of the output currents of said first and second means (10, 20), and fifth means (83) for producing an oscillating signal whose frequency is adjusted according to the integrated composite current value produced from said fourth means (30), said fifth means (83) comprising a voltage controlled oscillator whose one input terminal is supplied sequentially with first and second threshold voltages, one of the threshold voltages being dependent on the integrated composite current value produced from said fourth means (30), so that the oscillating frequency of said voltage controlled oscillator is dependent on the values of the first and second threshold voltages; characterized by sixth means (110) for frequency-dividing the oscillating signal; seventh means (120A) for synchronizing the frequency-divided signal with the oscillating signal, for controlling a pulse duration of the frequency-divided signal, and for supplying the synchronized and pulse-duration controlled signal to said first or second means (10, 20) as the first or the second signal, respectively; and eighth means (120B) for synchronizing the second signal with the oscillating signal, for controlling a pulse duration of the second signal, and for supplying the synchronized and pulse-duration controlled second signal to said second or first means (20, 10). An oscillator circuit according to claim 1, characterized in that said fourth means comprises (30) a differential amplifier (31) and a capacitor (32) for integration connected to the input and output terminals of said differential amplifier (31). An oscillator circuit according to claim 1 or 2, characterized in that said third means comprises: a DC voltage supply (VV1) for supplying a DC voltage to one of said first and second means (10, 20), and a variable DC power supply (VV2) for supplying a DC voltage to the other of said first and second means (10, 20) for adjusting the frequency of the oscillating signal produced from said fifth means (83). An oscillator circuit according to claim 1, characterized in that said sixth means (102) includes a frequency divider (110) for frequency-dividing the oscillating signal and for feeding the frequency-divided signal to said first or second means (10, 20) as the first signal.

Inventors: ISEKI, HIDEMI C/O PATENT DIVISION, SHIGEHARA, HIROSHI C/O PATENT DIVISION, YAMAGUCHI, AKIRA C/O PATENT DIVISION
Applicants: KABUSHIKI KAISHA TOSHIBA
Application year: 1984
Main classification: H03K3
IPC codes: H03K3, G01R23, H03K4, G10L19