METHOD OF AND APPARATUS FOR CODING AND DECODING PAL COLOUR TELEVISION SIGNALS

Abstract: A PAL coder operates to cause spectrum folding of a luminance input signal Y by multiplying twice the colour subcarrier (44) and adding (42), the resultant being comb filtered (F₁₂) to select frequencies which are multiples of the line frequency. The chrominance signals U, V are combined (48) into U + V and U - V on alternate lines and are then modulated onto subcarrier, the resultant being filtered in a comb filter (F₂₂ to select frequencies which are odd integral multiples of half the line frequency. The two signals are then added. The comb filters mentioned can in an alternative be based on 313 or 625 line delays. Prefiltering of the Y, U and V signals can be combined with the main comb filtering. The resultant circuitry can be arranged to minimise the use of delays, and uses analogue signals throughout. A converse decoder filters then spectrum folds the luminance component, and filters and multiplies by subcarrier to derive the chrominance. Once a television signal has been encoded by use of the encoder subsequent decoding and recoding produces no theoretical impairment into the signal.

Description: METhOD OF AND APPARATUS FOR CODING AND DECODING PAL COLOUR TELEVISION SIGNALS This invention relates to the coding and decoding of PAL colour television signals. Our British Patents 1,534,268 to 1,534,270, (Inventor: Martin Weston) of which No 1,534,269 may be taken as typical, describe a digital television transmission or processing system in which there are two transmitted signals, namely a luminance signal Y2 sampled at 2f , and a chrominance signal C1 sampled at f sc sc where f is the colour sub carrier frequency. Before sc digitisation the PAL chrominance signals U and V are combined into U t V on one television line and U - V on the next. The system can accept signals in separated chrominance and luminance form, i.e. YW form, or in encoded PAL form, and deliver output signals in YW or encoded PAL' form. Any of these signals may be in analogue form, or in digital form sampled at 4f . Each of the sc conversion stages of the system incoporates an appropriate comb filter which is based on the use of a line delay device. The transmitted signals Y2 and C1 form a package which is hereinafter referred to as W. By using the system, a country generating PAL signals can exchange programmes with a country generating YW signals using the common signal package W. This composite signal W is such that, if it was derived from PAL, it can be transformed back into a PAL signal with, in principle, no loss. This property of no (or substantially no) loss can be referred to as "transparency'1. A system is transparent if the system output is e essentially indistinguishable in practical terms from the system input (any overall delays being discsunted). The YUV output signals obtained from W, on the other hand, are impaired versions of the YW input signals (assuming these are vertically unfiltered). Our above men tioned patents mention that a PAL decoder can be constituted by using the PAL to W decoder connected to the circuits which convert the W components signals to Y and U, V respectively. We have also appreciated that the route YW-W-PAL' constitutes a PAL coder. Furthermore. as the route PAL-W-PAL' is transparent, it follows that in cascaded systems of the same type the route W-PAL' -PAL-W is also transparent. It can further be shown that in cascaded systems of the same type the route W-Y'U'V'-YUV-W is transparent. Thus it is possible, in princple, to cascade any number of systems of the same type, joined either by PAL or YUV ports without impairing the W signal package. In particular, the route PAL-W-Y'U'V'-YW-W-PAL is equivalent to the route PAL-W-PAL' which is, in principal, transparent. Thus, decoding using the system followed by recoding also using the system is transparent. Further the route YUV-W- PAl'-PAL-W-Y'U'V' is equivalent to the route Y W-W-Y'U'V', so that coding by the system followed by decoding by the system is equivalent to a YUV transmission path in which luminance is never mixed with chrominance. Thus it is possible to PAL code and decode using the system and eliminate luminance-chrominance cross-effects coi±rrlete' in exchange for the impairments introduced into the yW signals. These impairments are a reduction in vertical resolution of chrominance and vertical component resolution of high-frequency luminance, and the introduction of luminance aliasing and U-V crosstalk. Such coding and decoding is hereinafter referred to as clean coding and decoding. In accordance with another feature of this invention, this property of eliminating cross-effects depends only on the transparency of the W-PAL'-PAL-W path and thus is independent of the nature of the filters in the YUV-W and the W-Y'U'V' paths. The filters in these paths can thus be chosen with other factors in mind. In accordance with a further feature of the invention, the transparency of the W-PAL'-PALrW path is maintained if the delay element in the comb filters involved in this path is any odd number of lines, e.g. 313 or 625 lines. In accordance with yet another feature of this invention, the sampling in a PAL coder or decoder can be achieved by analogue multiplication followed by simple addition. Over the video band this provides spectrum folding in a manner identical to sub-Nyquist sampling. Preferably in constructing a PAL coder or decoder embodying the invention, the comb filters in the path between YW and PAL are collapsed so as to require a minimum of delay elements. The present invention is defined in the appended claims, to which reference should now be made. The invention will be described by way of example with reference to the drawings, in which: FIGURE 1 is a block diagram of a colour television transmission system based on our British Patent No 1,534,269; FIGURE 2 illustrates the form of the filters in the system of Figure 1; FIGURE 3 is a frequency diagram illustrating how analogue multiplication and addition are equivalent to sampling; FIGURE 4 is a block circuit diagram of a PAL coder embodying the invention; FIGURE 5 is a block circuit diagram of a PAL decoder embodying the invention; FIGURE 6 is a block circuit diagram of a rearrangement of the circuit of Figure 4; FIGURE 7 is a block circuit diagram of a rearrangement of the circuit of Figure 5; FIGURE 8 is a block circuit diagram of a further simplified rearrangement of the circuit of Figure 4; ; FIGURE 9 is a block circuit diagram of a further simplified rearrangement of the circuit of Figure 5; and FIGURE 10 is a block circuit diagram showing the form of the luminance stop filter used in Figures 8 and 9. In the system shown in Figure 1, signals can be received either in Y,U,V, form or in encoded PAL form, and can be outputted either in Y'U'Viform or encoded PAL' form. Various comb filters Fol, F31, F41, F11 and F21 are included at the input and comb filters F02, F32 F42 F12 and F22 are correspondingly included at the output as shown. Filters F11, F21 and F12, F22 are funeamente' to the system. Filters F11 and F12 provide a comb filter with a modulus sine response peaking at integral multiples of the television line frequency, while filters F and F22 provide a comb filter with a similar response but peaking at odd integral multiples of half the line frequency. The other filters, being those in the YUV-W and W-Y W paths and which we have now appreciated are not fundamental to the system, have responses which peak at multiples of the line frequency. It will be seen that over the regions where comb filtering takes place, the transmission/ frequency characteristic of all these filters is of a modulus sine form. In the central block marked W the signals comprise a luminance signal Y2 sampled at twice the colour subcarrier frequency (2f ) and a chrominance signal C1 sampled at the subcarrier sc frequency (f ) and consisting of U + V on one television line and U - V on the next. Switching circuits are accomodated in the V and V' signal paths to take account of the PAL switch. It will be seen that the input and output signals are in analogue form and hence analogue-to-digital converters (ADC's) and digital-toanalogue converters (DAC's) are included. As mentioned above, the elimination of cross-effects in the system depends only on the transparency of the W-PAL'-PAL-W path, and this is independent of the nature of the filters Fol, F31, and F41 in the YUV-W path and the filters F02, F32 and F42 in the W-Y'U'V' path. The nature of these filters does, however, govern the form of the impairments introduced into the YW signals. Figure 2 shows the form of all the filters except F01 and F02 as described in our Patent 1,534,269, filters F01 and F02 being identical to filters F11 and F12 respectivelly. For further details of these filters reference should be made to our British Patent 1,534,269. In Figure 2 of this application the same reference numerals are used as in our earlier patent. Now, the 2f sampling action in the luminance path from sc Y to W can be accomplished, over the video band, by analogue multiplication and addition. Referring to Figure 3, the uppermost waveform shows the effect of sampling the video baseband, itself shown in the second line, at a frequency such as 2f which is below sc the Nyquist limit, i.e. is less than twice the maximum video frequency. As is well known, the action of sampling causes the spectrum to be repeated infinitely at spacings equal to the sampling frequency Within the video band therefore, the sampled signal consists of the baseband signal together with a signal formed by spectrum folding at the upper end of the video band. The relation of these alias components to the television line frequency is well discussed in our earlier patents. The lowest line in Figure 3 shows what happens when the baseband signal is modulated onto a frequency of 2f c' or in other words is multiplied by it. Here the effect is not to cause an infinite number of repetitions of the video band, but rather to cause just two, at 2f c spacing from the baseband signal. We have sc appreciated that if this modulated signal is now added to the baseband signal, then over the video band the result is the same as is achieved bysampling, with spectrum folding occurring at the edges of the video band. The sampling action at f , low pass filtering and re-modulation in sc the chrominance path of the coder canbe accomplished by straightforward modulation. In the decoder7 sampling at f of the chrominance bandpass sc signal is equivalent to demodulation at the appropriate phase. Thus it is possible to construct an analogue coder and decoder which produce exactly the same signals as the digital system of our earlier Patents. Figures 4 and 5 show respectively an analogue PAL coder and decoder embodying the invention. In Figure 4, the input luminance, filtered by filter F011 is mixed in an adder 42 with a version of itself which is modulated in a modulator 44 on a carrier of frequency 2fsc. Such a modulator. as described above with reference to Figure 3, serves to introduce the luminance alias component, and may be termed a modifier. The mixture of wanted and alias luminance is then filtered by filter F12, which removes half-line frequency offset alias components, those at (n + ¸) fn , generated from line-offset input components, while allowing the components which are at multiples of the line frequency to pass. Finally, the signal is filtered by a low-pass filterLP2, which is the implicit luminance DAC filter in Figure 1, to remove components, particularly alias components, beyond the nominal video bandwidth. The U and V chrominance signals are, as in Figure 1, assumed to be limited in bandwidth to (h)f and are prefiltered by their so respective filters F31 and Fit1, the V signal having been PAL switched in a circuit 46. They are then combined in an adder 48 into a single chrominance signal in the form U + V and U - V on alternate lines, and modulated in a modulator 50 onto a PAL subcarrier (f ) whose phase is appropriately related to the sc modifier phase. This relationship is such that peaks of the subcarrier coincide with peaks of the twice-suboarrier (luminance) modifier frequency. The modulated chrominance is then passed through comb filter F22 which allows components which are at odd integral multiples of the line frequency to pass and restores the quadrature phasing of U and V signals in uniform-coloured areas. This modulated chrominance is then combined in an adder 52 with the luminance to give a PAL signal. In practice the video band filterLP2 can be placed after the adder 52. In the decoder of Figure 5, the PAL signal is split into two paths by filters F11 and F21. The effect of the filter F11, which averages across a one-line delay in the chrominance band and thus has peak response for line-offset components, is to co-phase the U and V modulated chrominance components in uniform-coloured areas. Thus by adding in an adder 62 a PAL-modifed version from multiplier 63 of the output of Fll to itself, the chrominance may be cancelled out, provided the modifier phase is appropriately chosen. The signal so processed is then filtered by F02 to reduce or eliminate, depending on Fol, the alias spectrum generated by the modifier. Finally, the filterIP2 ensures that the Y signal is confined to the nominal video bandwidth. The effect of filter F21, which takes half the difference across a one-line delay and thus has peak response for half line frequency offset components, is also to co-phase the U and V modulated chrominance and null the line-offset luminance. Thus the output of filter F21 requires only a single demodulation in demodulator 66 after which it is low-pass filtered by filter Ipl,which is the implicit chrominance DAC filter in Figure 1. Finally the combined chrominance signal is split into U and switched V components by fillters F32 and F42 and the V switch is removed by a circuit 68. We have now found that the delay elements in the filters in the W-PAL'-PAL-W path (the PAL filters) need not be one-line delays but can be of any odd number of lines. These filters are filters F , F12, F , and F22, and in particular they can be 11 12 21 22 313 or 625 lines long, i.e. essentially one field or one picture. 'Where the delay is N lines long, then the comb filter will have a modulus sine response with peaks at frequencies equal to fL being the line frequency. When longer delays are used, then the YUV output signals obtained by conventional decoding of the clean coder output may exhibit movement smearing. However, we have also appreciated that the other comb filters, namely the filters F01, F02, F31, F32, F41, and F42, are not as such fundamental to the coder/decoder operation although they can be chosen with a view to optimising system performance. Filters of interesed in the F01 and F02 positions are (a) where they take the form of F11 and F12 in figure 2 with a l or 313 or 625 line delay element, or (b) where they are preferably equal and of any one or two dimensional low pass form, cutting near the sub carrier frequency. Filters of special interestd in the F31, F32, F41 positions are (a) where they take the form of Figure 2 with a 1 or 313 or 625 line delay element, or (b) where they take weighted contributions from any number of consecutive lines in one field to form a vertical low pass filter cutting near 78 c/ph (cycles per picture height), such that F31 and F32 are equal, apart from a scale factor, and the weighting coefficients of F41 are those of F31 except that alternate values are reversed in sign. Thus, for example, it is possible to have filters based on one-line delays in the luminance path and on 313-line delays in the chrominance path, taking advantage of the lower bandwidth needed for chrominance in moving areas. If the group delays of the luminance and chrominance filters differ, then appropriate compensating delays must be inserted before or after the filters. The Table appended to this description lists combinations of luminance and chrominance filters of interest showing the delay element of the filters and the amount of compensating delay. The latter is distributed between the coder and decoder in such a way as to minimise the vertical-temporal misregistration at the output of the coder. In the Table, fifteen cases are given and for each the first line gives the magnitude of the main delay, and the second line (or second and third lines) gives the magnitude of the compensating delays. The letter n (possibly with subscripts) is used to mean an integral number of lines, while 1D and 2D mean respectively a filter of any one or two dimensional low pass form. A one dimensional filter is simply a low pass filter to give horizontal filtering, while a two dimensional filter includes line delays to give vertical filtering within a field. All the combinations of the Table could, in theory, be used with 313- or 625-line based filters at Fll, F121 F21 and F22 (the PAL filters). However, combinations of practical interest are: (a) all the combinations of the Table and line based PAL filters. (b) 313-line based combinations of the Table and 313-line based PAL filters. (c) 625-line based combinations of the Table and 625-line based PAL filters. It is possible that the combinations which include 625-line based filters would lead to unacceptable movement smearing, but the one (case 15) with 625-line filters in all positions corresponds to perfect transmission of stationary pictures, i.e. no crosseffects and no loss of resolution in the Y, U and V signals. For this case the clean coding produces an identical signal to conventional coding (discounting any delay), apart from reducing noise, and it is the clean decoder which is of primary interest. For moving pictures, an adaptive system could be devised which switched between 625-line and 313-line or 1-line based filters according to the decision of a motion sensor. With such an arrangement it would be necessary to ensure no difference of overall group delay between the different states. The luminance path in the coder of Figure 4 includes two comb filters in series, F01 and F12. These, and the other series connected comb filters, can be rearranged so that the circuits of Figures 4 and 5 become the circuits of Figures 6 and 7. In the circuit of Figure 6, the filters F1 and F2 control the wanted and alias luminance spectra respectively. Here again the technique of multiplication and addition is used to generate the required spectrum folding. In the chrominance path, it is possible to modulate the U and V filters onto sub carrier as shown and then to filter using filters F3 and F4 together with a multiplier 70 and adder 72 operating on the composite U,V signal. the filters F3 and F4 control the asymmethy of the quadrature-modulated chrominance spectra about their carriers. An equivalence exists between filters F1 and F2 and filters F01 and F12 and between Filters F3 and F4 and filers F31, F41 and F22, as discissed below. In the decoder of Figure 7 filters F5 and F6 control the wanted and alias luminance spectra respectively, whilst discriminating, with the aid of the modifier, against the subcarrier. Filters F7 and F8 control the wanted and cross-talk chrominance spectra, whilst discriminating against line-locked luminance. In the sane way as in the coder, an equivalence exists between filters F5 and F6 and filters F11 and Fo2, and between filters F7 and F8 and filters F22, F32 and F42. A problem with the arrangements of Figures 4 and 5 is that the filters cannot share storage because they are dispersed. This becomes significant if some of the filters are based on 313- or 625-line delay elements. However, the arrangements of Figures 6 and 7 permit storage sharing. The relationship between filters F1 and F8 and the filters of Figure 1 are given by: F1=Hc-F01F12 F2=-F01(F12+2+F12-2)H90* F3=-F22[(F31+1+F31-1)H45+(F41+1+F411)H45*] F4=-F22+2[F31+1H45+2-F41+1B45+2]+F22-2[31-1H45-2 F41-1H45-2] F5=Hd-F11F02 F6=F11 (F02+2+F02-2)H90* F7=-F21[(F32+F32-1)H45* (F42+1+F42-1)H45] F8=-F21[(F32+1F32-1)H45* -(F42+1+F42-1)H45] In these equations H and Hd are the transfer functions of c d the delays of the filters F01 F12 and F11F02 respectively, and 0 0 H45 and H90 are the transfer function of and 90 phase advance, an asterisk indicating the conjugate (phase retardation). ±n represents a transfer function shifted so that its origin lines at nf i.e. : :- - sc Fn(x) = F(x-nfsc) If the filters in the coder path of Figure 1 all use the same delay element i.e. are all based either on 1 or 313 or 625 line delays, then filters F3 and F4 are respectively equal to filters F1 and F2. Similarly if the filters in the decoder path of Figure 1 satisfy the same condition, then filters F7 and F8 are respectively equal to filters F5 and F6. Then the coder and decoder can be further simplified to the arrangements of Figures 8 and 9, where the filters L1 and L2 are luminance-stop filters. In the coder the luminance-stop filter L is used to pass the modulated chrominance, thus making the luminance and chrominance spectra complementary. In the decoder the filter L separates out the modulated chrominance for quadrature demodulation. If the system of Figure 1 uses the one- line-b ased filters of Figure 2, Where F01 and F02 are respectively equal to F11 and F12, then the form of L is the same in both coder and decoder and is shown in Figure 10. The decoder compensating delay may be part of L as shown. The luminance stop filter L shown in Figure 10 will first be described and is seen to consist of hvo one-line delays 150, 152 in series, the output of delay 150 constituting an auxiliary output used in the decoder. The average of the input and two-line-delayed signals is taken in a halving adder 154 and this is subtracted from the one-line delayed signal in a subtractor 156. The difference between the input and two-line-delayed signals is also taken in a halving subtractor 158, and the phase of this signal retarded by 90 degrees in a circuit 160. The resultant is multiplied in a multiplier 162 by twice subcarrier at appropriate phase, and the resultant is added to the output of subtractor 156 in an adder 164. The adder output is band pass filtered in a chrominance filter 166. This filter L is used in both Figures 8 and 9. The filter is seen to provide essentially all the functions of the filters F1 to F4 in Figure 6 and of F5 to F8 in Figure 7. This filter is therefore particularly advantageous in reducing the number of delay elements required. More generally, the same arrangements are applicable if 313- or 625-line based filters are used throughout, when the delay elements of Figure 10 are modified accordingly. However, in the 313-line case the transfer functions H and H represent 45 90 phase shifts of -45 ans -90 respectively and the demodulation phase of U is reversed. Any of the circuit arrangements of Figures 4 to 10 may be realised digitally using an arbitrary sampling frequency in the region of three times the subcarrier frequency. However, a bandpass filter must then precede the modifier wherever it occurs, as shown dotted in Figure 10. The lower cut-off frequency of the bandpass filter must lie between f - 2if and 2f - 5if ,where f is the s sc s sc 5 digital sampling frequency. The upper cut-off frequency must lie between the nominal video bandwidth ana half the sampling frequency. Table Delay basis in lines of luminance and chrominance filters used in Figure 1 and compensating filters EMI16.1 <tb> Case <SEP> F01 <SEP> F02 <SEP> F31/F41 <SEP> F32/F42 <tb> 1 <SEP> 1D <SEP> 1D <SEP> 2n <SEP> 2n <tb> <SEP> n <SEP> n <SEP> 0 <SEP> 0 <tb> 2 <SEP> 1D <SEP> 1D <SEP> 2n+1 <SEP> 2n+1 <tb> <SEP> n+1 <SEP> n <SEP> 0 <SEP> 0 <tb> 3 <SEP> 1 <SEP> 1 <SEP> 2n+1 <SEP> 2n+1 <tb> <SEP> n <SEP> n <SEP> 0 <SEP> 0 <tb> 4 <SEP> 2D <SEP> (group <SEP> delay <SEP> n1) <SEP> 2n2 <SEP> 2n2 <tb> <SEP> n2 > n1 <SEP> n2-n1 <SEP> n2-n1 <SEP> 0 <SEP> 0 <tb> <SEP> n2 < n1 <SEP> 1 <SEP> 0 <SEP> n1-n2 <SEP> n1-n2 <tb> 5 <SEP> 2D <SEP> (group <SEP> delay <SEP> n1) <SEP> 2n2+1 <SEP> 2n2+1 <tb> <SEP> n2 > n1 <SEP> n2-n1+1 <SEP> n2-n1 <SEP> 0 <SEP> 0 <tb> <SEP> n2 < n1 <SEP> 1 <SEP> 0 <SEP> n1-n2 <SEP> n1-n2 <tb> 6 <SEP> 1D <SEP> 1D <SEP> 313 <SEP> 313 <tb> <SEP> 313 <SEP> 0 <SEP> 0 <SEP> 0 <tb> 7 <SEP> 1 <SEP> - <SEP> 313 <SEP> 313 <tb> <SEP> 312 <SEP> 0 <SEP> 0 <SEP> 0 <tb> 8 <SEP> 2D <SEP> (group <SEP> delay <SEP> n) <SEP> 313 <SEP> 313 <tb> <SEP> 313 <SEP> 0 <SEP> n <SEP> n <tb> 9 <SEP> 313 <SEP> 313 <SEP> 313 <SEP> 313 <tb> <SEP> 0 <SEP> 0 <SEP> 0 <SEP> 0 <tb> 10 <SEP> 625 <SEP> 625 <SEP> 313 <SEP> 313 <tb> <SEP> 0 <SEP> 0 <SEP> 312 <SEP> 0 <tb> 11 <SEP> 1D <SEP> 1D <SEP> 625 <SEP> 625 <tb> <SEP> 313 <SEP> n312 <SEP> 0 <SEP> 0 <tb> 12 <SEP> 1 <SEP> 1 <SEP> 625 <SEP> 625 <tb> <SEP> 312 <SEP> 312 <SEP> 0 <SEP> 0 <tb> 13 <SEP> 2D <SEP> (group <SEP> delay <SEP> n) <SEP> 125 <SEP> 625 <tb> <SEP> 313-n <SEP> 312-n <SEP> 0 <tb> 14 <SEP> 313 <SEP> 313 <SEP> 625 <SEP> 625 <tb> <SEP> 0 <SEP> 312 <SEP> 0 <SEP> 0 <tb> 15 <SEP> 625 <SEP> 625 <SEP> 625 <SEP> 625 <tb> <SEP> 0 <SEP> 0 <SEP> 0 <SEP> 0 <tb>

Claims: CLAIMS 1. A method of encoding Y, U and V signals into a PAL colour television signal, comprising: spectrum folding the luminance signal Y with respect to a frequency twice the colour sub carrier frequency to produce alias components in the video band; comb filtering the resultant luminance signalwith amodulus sine response having peaks at integral multiples of fL/N, where f is L L the television line frequency, and N is an odd integer; forming the U and V signals into a composite chrominance signal C based on U + V and U - V on alternate lines; modulating the chrominance signal C onto the colour subcarrier frequency; comb filtering the modulated chrominance signal with a modulus sine response having peaks at odd integral multiples of fT/2N; ; and combining the luminance and modulated chrominance signals. 2. A method according to claim 1, including additional filtering to prefilter the Y, U and V signals with a comb filter response. 3. A method of decoding a PAL colour television signal into Y, L7 and V signals, comprising: comb filtering the PAL signal with a modulus sine respcnse having peaks at integral multiples of f /N, where f is L L the television line frequency, and N is an odd integer; spectrum folding the said comb filtered signal with respect to a frequency twice the colour subcarrier frequency to provide a l1rinance signal; comb filtering the PAL signal with a modulus sine response having peaks at odd integral multiples of f /2N; L multiplying the last-mentioned comb filtered signal by a signal of subcarrier frequency; and separating the multipled signal into its orthogonal U and V components. 4. A method according to claim 3, including additional filtering to post-filter the Y, U and V signals with a comb filter response. 5. A method according to claim 2 or 4, in which the additional filtering steps are combined with the aforesaid comb filtering steps respectively. 6. A method according to any preceding claim, in which N is greater than one, and is preferably 313 or 625. 7. A method according to any preceding claims, in which the spectrum folding is achieved by multiplication by a signal twice the sub carrier frequency and addition of the resultant to the original signal. 8. A method according to any preceding claim, in which the luminance and chrominance signals remain in analogue form. 9. A method of decoding in accordance with claim 3 sequentially followed by a method of encoding The decoded signals in accordance with claim 1. 10. A PAL coder for encoding Y, U and V signals into a PAL colour television signal, co:rnrisThg: luminance processing circuitry adapted to cause spectrum folding of a luminance input signal Y with respect to a frequency twice the colour sub carrier frequency to produce alias components in the video band, and to como filter the resultant luminance signal with a modulus sine response having peaks at integral multiples of f /N, where f is the television line frequency, and N is an L L odd integer; ; chrominance processing circuitry adapted to form the U and V signals into a composite chrominance signal C based on U + V and U - V on alternate lines, to modulate the chrominance signal C onto the colour subcarrier frequency, and to comb filter the modulated chrominance signal with a modulus sine response having peaks at odd integral multiples of fL/2N; and combining means for combining the luminance and modulated chrominance signals. 11. Apparatus according to claim 10, including additional filters for prefiltering the Y, U and V signals with a comb filter response. 12. . PAL decoder for decoding a PAL colour television signal into Y, U and V signals, comprising. luminance processing circuitry adapted to ccmb filter the PAL signal with a modulus sine response having peaks at integral multiples of fL/N, where fL is the television line frequency, and N is an odd integer, and to cause spectrum foldin of the com filtered signal with respect to a frequency twice the colour subcarrier frequency to provide a luminance signal; chromginance processing circuitry adapted to comb filter me PAL signal with a modulus sine response having peaks at odd integral multiples of fL/2N, to multiply the last-mentioned comb filtered signal by a signal of subcarrier frequency, an to separate the multiplied signal into its orthogonal U and V components. 13. Apparatus according to claim 12, including additional filters for post-filtering the Y, U and V signals with a comb filter response. 14. Apparatus according to claim 11 or 13, in which the additional filters are combined with the aforesaid comb filters respectively. 15. Apparatus according to any of claims 10 to 14, in which N is greater than one, and is preferably 313 or 625. 16. Apparatus according to any of claims 10 to 15, in which the spectrum folding ls achieved by multiplication b a signal of twice the subcarrier frequency and addition of the resultant to the original signal. 17. Apparatus according to any of claims 10 to 16, in which the processing circuitry operates with the signals remaining in analogue form. 18. A decoder according to claim 12 having an encoder according to claim 10 coupled to its outputs. 19. A PAL coder comprising first, second and third inputs for receiving Y, U and V input signals respectively; V-axis switching means for switching the V signal; multipliers for modulating the U and V signals onto subcarrier at orthogonal phase positions; combining means for combining the U and V signals and subtracting them from the Y input signal; a luminance-stop comb filter connected to the output of the combining means; and a sub tractor for subtracting the thus filtered signal from the luminance input signal to provide a PAL output signal. 20. A PAL decoder comprising an input for receiving a PAL input signal; a luminance-stop comb filter connected to the input; a subtractor connected to subtract the output of the luminance-stop filter from the input signal to provide a Y output signal; multipliers for multiplying the output of the luminance-stop filter by two subcarrier signals of orthogonal phase to provide a U output signal and a switched V signal; and V-axis switching means for removing the V-axis switching to provide a V output signal. 21. Apparatus according to claim 19 or 20, in 'nich the luminance-stop filter oomprises first and second delays connected in series each of an odd number of lines, combining means for averaging the input and the output of the second delay and subtracting the resultant from the output of the first delay; half-differencing means for taking half the difference between the input and the output of the second delay; multiplying means for multiplying the output of the half-differencing means by a signal of twice sub carrier frequency; and an adder for adding the outputs of the combining means and the multiplying means.

Inventors: CLARKE, CHRISTOPHER KEITH PERRY, DREWERY, JOHN OLIVER, WESTON, MARTIN
Applicants: BRITISH BROADCASTING CORPORATION
Application year: 1980
Main classification: H04N11
IPC codes: H04N11