# Patent

## Title
Method and apparatus for driving half bridge connected semiconductor power switches with a stable and extremely short interlock delay combined with a switching transition speed increase and a driving power consumption reduction

## Abstract

A driving circuit for driving half bridge connected electrically controlled power switches with a near zero interlock delay time between on-states of the power switches, wherein the driving circuit is configured to receive an input signal and to generate: —a first drive signal being adapted to switch a first power switch between the on and off state, —a second drive signal being adapted to switch a second power switch between the on and off state, wherein the signal curve of the first drive signal generated in response to a rising and falling edge of said input signal is mirrored with respect to the signal curve of the second drive signal along a time axis of a mirroring voltage value within a transition time, wherein the mirroring voltage value is adjusted such to be within the cutoff region of the power switches.

## Background
TECHNICAL BACKGROUND
The invention relates to a method and a circuit for driving semiconductor power switches, such as Bipolar Junction Transistor (BJT), Metal Oxide Silicium Field Effect Transistor (MOSFET), Silicium Carbide Field Effect Transistor (SiC FET), Galium Nitride Field Effect Transistor (GaN FET) or Junction Field Effect Transistor (JFET) transistors connected in a half bridge HB configuration, which are mainly used as a switch unit to convert power from one form to another in electric motor and generator controlled systems, uninterrupted and interrupted power supplies, voltage DC transmission and in many other conversion applications. Half bridge connection of the power switches controlled with pulse width modulation (PWM) or pulse density modulation (PDM) is a commonly used technique for these applications.
In most conversion applications the half bridge connected power switches switch between both poles of the applied voltage. By alternating the ratio of the conducting time of both power switches and simultaneously setting the duty cycle, the generated output voltage value is defined. The advantage of such a power conversion enables the bidirectional conversion between DC-DC, DC-AC or AC-AC voltage levels theoretically without losses.
Both power switches alternate among two states, namely an on-state and off-state. During the off-state the resistance of the power switch is very high, therefore no load current can flow through it and further no losses are generated, while during the on-state the power switch conducts the load current and the conductive losses are generated due to the internal on-resistance. The value of the internal on-resistance is low enough for the conductive losses to be very low. Changing the states of the individual power switch causes very high losses due to the simultaneity of the operating voltage and the entire load current. Speeding up the state changes reduces the so called switching losses. For the transient time duration the turning-on power switches to over take the entire load current, which in case of the so called hard switch includes also the body diode charge of the second power switch. Accordingly, during a hard switch the switching losses and electro-magnetic interferences EMI increase drastically.
Only one of the two in half bridge connected power switches may be switched on at the same time, since the cross conduction current between the power supply terminals over the power switches would cause a permanent damage to the power switches. In order to prevent any possibility that such a situation can occur the command to turn-on off the second power switch must be performed with an additional minor delay, so-called dead time DT, after the command to turn-off the first power switch. The minimum value of the dead time DT is given by a worst case combination of the driver signal propagation delay and transition time, as well as propagation delay and rise time of the power switches under all possible working conditions, such that a certain time delay, namely interlock delay ID, exists between the executed turn-off of the first power switch and the beginning of the turning-on of the second power switch. The voltage switch between both power switches can be done either by a load inductivity, which constitutes the so called soft switching or by one of the power switches, which constitutes the so called hard switching. The switching type, whether it is a soft or hard switching depends on the load current direction.
Increasing the dead time DT also increases the reliability of the circuit against cross conduction currents, however a longer interlock delay diminishes a switching ratio especially for high switching frequencies and causes distortion of the transmitted signal. During the time of the interlock delay the load current flows via the body diode of one of the two power switches causing additional losses due to worse switching characteristics of the body diode in comparison with the turned-on power switch. A further aspect of a long interlock delay are the charge carriers, which occur in the body diode conducting time. These charge carriers, which form the so called reverse recovery charge in the first power switch are discharged by the second power switch. Such situation is known as hard switching.
A conventional way of reducing such switching losses is by adding an additional resonant circuit allowing a soft switching. In the case that the hard switching is not eliminated the common way of reducing the switching losses is the reduction of the PWM frequency or by providing a circuit having an additional fast diode connected in parallel with the source S and drain D of the power switches PT1 and PT2, which over takes the load instead of the parasitic diode. The reduction of a working PWM frequency can be only useful for the application circuits having a sufficient inductance of the load.
An ideal hard switching with minimum power losses could theoretically be reached in the case of an interlock delay short enough (within 10 nsec range), such that the charge carriers do not form and further no reverse recovery in the hard switch situation can occur. Therefore, the switching losses would be significantly smaller.
In order to follow the trend of decreasing the weight and size of the switching conversion devices increasing the carrier frequencies obligatory. In this way a higher resolution and higher frequencies of the generated output voltage can be achieved. With the switching frequency increase the response time for rapid load changes becomes shorter and operation frequencies high enough even for the modern electro motors and audio class D amplifiers will be achieved.
The state-of-art power switch driving technologies do not reduce the interlock delay such that it becomes short enough to decrease the switching losses to a range which allows to significantly increase the switching frequencies in order to follow the later trend.
STATE OF THE ART
There are two known ways of preventing a simultaneous switching on of power switches connected in a half bridge, wherein the first way is demonstrated in FIG. 24, and the second way is shown in FIG. 26. The FIGS. 25 and 27 show the corresponding time curves.
In the conventional driving circuit for driving half bridge connected power switches the logic circuit Log C shown in FIG. 24 converts the input signal A to driving signals B1 and C1 having the rising edges delayed by the delay unit Del for a pre-set delay time DT. The value of the pre-set dead time DT has to take into account the worst case situation delays of the control lines and the worst case situation delays of the power switches PT1 and PT2 to ensure a positive value of the interlock delay ID between the on-states of the power switches PT1, PT2 during the operation, such that the cross conduction cannot occur. The first output signal B1 of the logic circuit Log C is amplified through a first galvanically isolated driving circuit DR1 and converted by means of a resistor RG1 into the signal P1, which switches among the floating supply voltages Vs+′,Vs−′ output by the a first power supply DC′ and controls the on- and off-states of a first power switch PT1. The second output signal B2 of the logic circuit Log C is amplified through a second driving circuit DR2 and converted by means of a resistor RG2 into the signal P2, which switches among the supply voltages Vs+, Vs− out put by the power supply DC′ and controls the on- and off-states of a second power switch PT2. The resistors RG1 and RG2 define the voltage slope and accordingly the speed of the switch execution of the power switches PT1 and PT2. The power switches PT1, PT2 of the circuit as shown in FIG. 24 are formed by power MOSFETs.
FIG. 25 shows the signal diagrams of the signals for the conventional driving circuit according to the state of the art as shown in FIG. 24. The input PWM signal A and output signals B1 and B2 of the logic circuit Log C are demonstrated with logic levels 0 and 1. The signals B1 and B2 are mutually inverted and delayed for the logic propagation delay tlpd with respect to the input PWM signal A. Additionally, each rising edge of the signals B2 and B2 is delayed for the preset dead time DT generated within the delay unit Del. The signals P1 and P2 are delayed for a driving propagation delay tdpd of the driving circuits DR1 and DR2 with respect to the corresponding signals B2 and B2. In FIG. 25 the signals P1 and P2 are shown for the case that the supply voltages Vs− and Vs−′ is connected to the source S of the respective transistor and has a value of 0V, thus the positive voltage control pulses that drive the power switches PT1 and PT2. During the time when the signal P1 is higher than the threshold voltage Vth of the first power switch PT1 the first power switch PT1 is on and similarly the second power switch PT2 is on during the time when the threshold voltage Vth of the second power switch PT2 is exceeded by the value of the signal P2. The time when none of the two power switches PT1 or PT2 is on is called the interlock delay ID. The variations in the logic propagation delay tlpd, the driving propagation delay tdpd and the threshold voltages Vth1, Vtw2 of the power switches PT1, PT2 cause a variation of the interlock delay ID. In order to prevent a negative interlock delay ID causing the undesired cross connection the preset dead time DT needs to be long enough and has to take into account worst case situations. To prevent a cross connection situation the state of the art driving circuits use preset dead times DT, which is accordingly to the stated above limited down to at least 50 nsec and mostly over 100 nsec.
FIG. 26 shows an alternative implementation of a driving circuit according to the state of the art without the delay unit Del such that the signal B1′ and B2′ are mutually inverted and amplified via isolated driving circuit DR1 and driving circuit DR2. The summation of the propagation delays tlpd and tdpd of both driving lines are adjusted to be equal. The time difference for a delayed turn-on with respect to the turn-off is defined by the difference of the RC′, RC′′ time constant formed within the circuits RC1 and RC2 when loading (RC′) or disloading (RC′′) the gate-source capacitors of the power switches PT1 and PT2. The driving implementation illustrated in FIG. 26 and the corresponding time curves in FIG. 27 show a cost effective solution, which is useful for low power and low frequency applications with narrower duty cycle range. Since a long turn-on RC′ constant causes a slower on switching of the power switches PT1 and PT2 the switching losses are significantly higher. The shown conventional solution often uses the powering via the bootstrap diode Dbs as shown in FIG. 26.
The state of the art solution shows several limitations regarding improvements of the switching performance of power switches PT. Consequently the use of higher operating frequencies is limited due to the drastic increase of the switching losses with increasing frequency.
SUMMARY OF THE INVENTION
Accordingly, it is an object of the present invention to provide a method and a circuit for driving half bridge connected semiconductor power switches with a stable and extremely low interlock delay combined with switching transition speed increase and driving power consumption reduction.
This object is achieved by a driving circuit for at least one half bridge connected power switches comprising the features of claim 1.
Accordingly the invention provides a driving circuit for driving half bridge connected electrically controlled power switches with a near zero interlock delay time between on-states of the power switches, wherein the driving circuit is configured to receive an input signal and to generate two driving signals as a response to the rising and falling edge of the input signal, wherein each driving signal is adapted to switch the state of a corresponding power switch on and off and the time curves of the aforementioned driving signals are mirrored with respect to one another along a time axis through a mirroring voltage value within the transition time, wherein the mirroring voltage value is adjusted to be low enough for the power switches to be within the cutoff region.
In a possible embodiment the applied input signal is a pulse width modulated (PWM) signal.
In a possible embodiment the applied input signal is a pulse density modulated (PDM) signal.
In a possible embodiment the driving circuit according to the present invention comprises at least one triggering circuit and at least one ramp generating circuit for controlling each power switch. Each triggering circuit comprises two triggering switches adapted to generate a triggering signal. A first triggering switch fulfills the function of switching the off-state voltage value to the positive DC gate supply voltage value, while the second triggering switch fulfills the function of switching the on-state voltage value to the negative DC gate supply voltage value. The triggering switches can be any electrically controlled switching element or any combination of electrically controlled switching elements. Every even change of the input signal simultaneously triggers the first triggering switch that controls the first power switch and the second triggering switch that controls the second power switch. Every odd change of the input signal simultaneously triggers the second triggering switch that controls the first power switch and the first triggering switch controls the second power switch. The triggering signals of the power switches are further shaped into a drive signal via a ramp generating circuit by forming a slope. Each power switch comprises at least one or any combination of electrically controlled semiconductor switching elements. A switching element or a group of switching elements can be formed for example by a Si MOSFETs, a SiC MOSFETs, a GaN MOSFETs, a SiC JFETs or a SiC bipolar transistors, etc. Every switching element of first power switch is controlled by the first drive signal and every switching element of second power switch is controlled by the second drive signal.
In a possible embodiment of the driving circuit according to the present invention the driving circuit comprises at least one ramp generating circuit for each power switch comprising an energy buffer component formed by an inductivity connected between a common point of triggering switches of the corresponding triggering circuit and the control electrode of the corresponding power switch. The inductivity is adapted to supply a maximum current to the control electrode of a corresponding power switch at the time when the threshold voltage of said power switch is reached. The driving energy stored in said inductivity within the switch time ts accelerates the switching of said power switch.
In a possible embodiment of the driving circuit according to the present invention a damping resistor adapted to damp a signal sway of the driving signal is connected in parallel to the inductivity.
In a possible embodiment of the driving circuit according to the present invention the driving circuit comprises a gate supply switching circuit and a power supply magnetic circuit. The triggering circuit additionally comprises a DC gate supply circuit. The gate supply switching circuit is configured to:

    
    
        receive a gate power ratio control signal,
        receive a DC gate power source voltage,
        generate a signal adapted to carry a gate supply energy and
        adjust the voltage value of the DC gate supply voltages out of the said gate supply circuits by modulating the said DC gate power source voltage with the PWM information of the gate power ratio control signal.
    
    


The power supply magnetic circuit comprises at least one transformer and is adapted to ensure a galvanically isolated energy transfer of the gate supply energy from the gate supply switching circuit to the DC gate supply circuits. The DC gate supply circuit comprises at least one half wave rectifier and at least one smoothing capacitor and is adapted to:

    
    
        generate one, two or three DC gate supply voltages.
        supply the triggering switches and
        receive energy from a power supply magnetic circuit.
    
    


Each DC gate supply voltage can have a value of 0V.
In a possible embodiment of the driving circuit according to the present invention the driving circuit comprises a temperature influence adapting circuit adapted to:

    
    
        receive a temperature information signal,
        generate a gate power ratio control signal,
        set the value of DC gate power source voltage via the DC/DC converter and to
        set the gate power ratio control signal and the DC gate power source voltage with respect to the temperature information such that the value of DC gate supply voltages preserves a constant value of differential voltages over an entire temperature range of temperature information signal.
    
    


In a possible embodiment of the driving circuit according to the present invention the magnetic coupling circuit being connected between the at least one ramp generating circuit of the first power switch and the at least one ramp generating circuit of the second power switch is adapted to perform the mirroring between the trigger signals. The magnetic coupling circuit is formed by at least one transformer and is adapted to perform a coupling ratio 1 between the ramp generating circuits of the same power switch and a cooling ratio −1 between the ramp generating circuits of different power switches.
In a possible embodiment of the driving circuit according to the present invention the driving circuit comprises a tap on at least one winding of the magnetic coupling circuit and at least one triggering circuit comprising capping diodes connected to said tap. This additional tap forms an autotransformer function with a coupling ratio between 0.5 and 1. The magnetic coupling circuit is adapted for the mentioned coupling ratio to define a maximum voltage oversway value on the control electrode of the power switch. The magnetic coupling circuit and the capping diodes are adapted to return at least part of the driving energy back to the DC gate supply circuits via the capping diodes.
In a possible embodiment of the driving circuit according to the present invention the driving circuit comprises a differentiating circuit and a magnetic triggering circuit. The differentiating circuit is configured to receive an input signal and to generate pulsed signals. The pulsed signals switch among an active and inactive state and are adapted to control the triggering switches through the magnetic triggering circuit without any additional amplification. The value of the first pulsed signal is during the on state of the input signal set to an active state for a preset time duration at least in response to each rising edge of the input signal. The value of second pulsed signal is during the off state of the input signal set to an active state for the preset time duration at least in response to each falling edge of said input signal. The preset time has either a predetermined value or it is defined simultaneously with respect to a power switch state information in order to ensure a minimal driving power consumption. An additional winding of the magnetic coupling circuit is adapted to generate power switch state information. The magnetic triggering circuit comprises at least one transformer and is adapted to:

    
    
        turn on the first triggering switch that controls the first power switch and the second triggering switch that controls the second power switch in response to every odd state change of first pulsed signal,
        turn on the second triggering switch that controls the first power switch and the first triggering switch that controls the second power switch in response to every even state change of first pulsed signal and
        hold the all triggering switches off in response to none of the pulsed signals being in an active state.
    
    


In a possible embodiment of the driving circuit according to the present invention the driving circuit comprises at least two bistable circuits adapted to prevent the changes in the conducting state of the power switches for the time when none of the triggering switches is switched on.
In a possible embodiment of the driving circuit according to the present invention the driving circuit comprises at least one shut down circuit adapted to switch off the power switches in response to shut down information received through the signal from the magnetic triggering circuit.
In a possible embodiment of the driving circuit according to the present invention the at least one triggering circuit comprises a diode. The aforementioned diode can be either connected between the DC gate supply circuit and the second triggering switch or between the DC gate supply circuit and the first triggering switch.
In a possible embodiment of the driving circuit according to the present invention the differentiating circuit and the magnetic triggering circuit drive at least four power switches connected into a full bridge within the same driving circuit.



BRIEF DESCRIPTION OF FIGURES
In the following possible embodiments of the method and circuit for driving at least one voltage controlled power switch are described with reference to the enclosed figures.
FIG. 1 shows a block diagram of a possible embodiment of a driving circuit for a voltage controlled power switch according to the present invention;
FIG. 2 shows the mirroring function of the driving circuit;
FIG. 3 shows a flow chart of a possible embodiment of a method for driving a voltage controlled power switch according to the present invention;
FIG. 4 shows a circuit diagram of a possible embodiment of a driving circuit according to the present invention;
FIG. 5 shows signal diagrams for illustrating the operation of a differentiating circuit according to the present invention;
FIG. 6 shows signal diagrams for illustrating the operation of a ramp generating circuit according to the present invention;
FIG. 7 shows signal diagrams for illustrating reduction of driving energy consumption of the driving circuit according to the present invention;
FIG. 8 shows a circuit diagram of a possible embodiment of powering of the driving circuit according to the present invention;
FIG. 9 show signal diagrams for illustrating the operation of the powering of the driving circuit according to the present invention;
FIG. 10 shows signal diagrams for illustrating the interlock delay dependency according to the present invention;
FIG. 11 shows a circuit diagram of a possible embodiment of temperature influence adapting circuit according to the present invention;
FIG. 12 show signal diagrams for illustrating the operation of temperature influence adapting circuit according to the present invention;
FIGS. 13a,b,c,d shows a current flow for the energy returning during the on switching of the power switch;
FIGS. 14a,b,c,d shows a current flow for the energy returning during the off switching of the power switch;
FIG. 15 shows a circuit diagram of a possible embodiment of driving circuit according to the present invention for driving a BJT;
FIG. 16 shows a circuit diagram of a possible embodiment of driving circuit according to the present invention for driving a normally off JFET;
FIG. 17 show signal diagrams for illustrating the operation of driving circuits according to the present invention shown on FIGS. 14-15;
FIG. 18 show signal diagrams for illustrating the operation of driving circuits according to the present invention shown on FIGS. 14-15;
FIG. 19 shows a circuit diagram of a possible embodiment of driving circuit according to the present invention for driving a normally on JFET;
FIG. 20 shows a circuit diagram of a possible embodiment of driving circuit according to the present invention for driving a SiC FET;
FIG. 21 show signal diagrams for illustrating the operation of driving circuits according to the present invention shown on FIG. 19;
FIG. 22 shows a circuit diagram of serial connected power switches according to the present invention;
FIG. 23 shows a circuit diagram of the driving circuit according to the present invention for driving two half bridges;
FIG. 24 shows a state of art driver;
FIG. 25 shows a state of art driver;
FIG. 26 shows a state of art driver;
FIG. 27 shows a state of art driver;



DETAILED DESCRIPTION OF FIGURES
As can be seen in FIG. 1 a driving circuit 1 according to the present invention is provided for driving half bridge connected electrically controlled power switches PT1, PT2 in terms of having a near zero interlock delay (NZID) time between on-states of the power switches PT1, PT2. The interlock delay ID is known as the time interval, which is necessary between switching the load from a first power switch PT1 to a second power switch PT2 and back from the second power switch PT2 to the first power switch PT1 in order to prevent a cross-conduction through the power switches PT1 and PT2. During the interlock delay ID time a body diode DB conducts a load current IL. The body diode DB is either a parasitic diode of a power switch PT1,PT2 or a diode connected in parallel of the respective power switch PT1 or PT2. The body diode DB conducts in a source drain direction of the power switch PT1 or PT2. The load current IL causes charge carriers to develop within the body diode DB. These charge carriers cause unwanted current spikes when one power switch PT1 or PT2 starts to overtake the load current IL from the body diode DB of the other power switch PT2 or PT1. An interlock delay ID under 100 nsec influences the amount of the charge carriers developed in the body diode DB. The lower the interlock delay ID is, the lower is the amount of the developed charge carriers, since there is no time for them to occur. Below an interlock delay ID value of about 10 nsec to 15 nsec the charge carriers do not appear on the body diode DB. The term near zero interlock (NZID) refers to an interlock delay, which is short enough for the charge carriers in the body diode DB not to occur.
A load can be connected to the output of the half bridge HB. The half bridge HB is formed by the first power switch PT1 and the second power switch PT2. The power switch PT1 and PT2 can be one switching element PT1a, PT1b . . . PT1n, PT2a, PT2b . . . PT2n or a plurality of aforementioned switching elements connected in an arbitrary interconnection, whether parallel, serial or a combination of both. The switching element PT1a, PT1b . . . PT1n, PT2a, PT2b . . . PT2n can be any electrically controlled switching element on a semiconductor basis such as Si, SiC, GaN, . . . as for example MOSFET, BJT, JFET, or any combination of these as for instance IGBT. As shown in FIG. 4 each power switch PT1 and PT2 consists of a FET cell FC and the aforementioned body diode DB. In case the body diode is not integrated into the respective power switch PT1, PT2 an additional diode is connected in parallel to each power switch PT1, PT2, such that the body diode DB conducts in the source to drain direction as shown in FIG. 12.
The driving circuit 1 shown in FIG. 1 receives an input signal A. The input signal A can for example be a pulse width modulated (PWM) signal or a pulse density modulated (PDM) signal. The driving circuit 1 generates at least one first drive signal J1 and at least one second drive signal J2, that switches between on and off states in response to every rising and falling edge of the input signal A. In the embodiment illustrated in FIG. 1 the first drive signals J1 controls the first power switch PT1 and the second drive signal J2 controls the second power switch PT2. Each switching element PT1a, PT1b . . . PT1n that forms the first power switch PT1 is driven with the drive signals J1 and each switching element PT2a, PT2b . . . PT2n that forms the power switch PT2 is driven with the second drive signals J2. The on state value of the drive signals J1 and J2 holds the corresponding power switch PT1 or PT2 on. The off state value of the drive signals J1 and J2 holds the corresponding power switch PT1 or PT2 off.
The functionality of the driving circuit 1 is demonstrated by exemplary time curves of an example of a soft and hard switch of a half bridge HB made of Si MOSFET power switches PT1 and PT2 in FIG. 2, wherein the switch waveforms on the left show a soft switching and the switch waveforms on the right show a hard switching. The soft switching occurs when the load current IL is transferred from the first power switch PT1 to the second power switch PT2 and hard switching occurs when the load current IL is transferred from the second power switch PT2 to the first power switch PT1. In the possible embodiment of FIG. 2 the first drive signal J1 controls the corresponding first power switch PT1 and the second drive signal J2 controls the corresponding second power switch PT2.
In the embodiment of the FIG. 2 the positive DC gate supply voltage VG+ represents the on state value of the drive signals J1 and J2. For the time of the on state value of the drive signals J1 and J2 the corresponding power switch PT1, PT2 is conducting deeply in the ohmic region and is therefore switched on. The negative DC gate supply voltage VG− represents the off state value of the drive signals J1 and J2. For the time of the off state value of the drive signals J1 and J2 the corresponding power switch PT1, PT2 is deep in the cutoff region and is therefore switched off. The cutoff region represents the voltage range of a voltage applied to the control electrode of the power switch PT, which ensures that the respective power switch PT is not conducting.
During the high value of the input signal A the on state value of the first drive signal J1 holds the first power switch PT1 on and the off state value of the second drive signal J2 holds the second power switch PT2 off. In the embodiment according to the FIG. 2 the on state value of drive signals J1 and J2 is the positive DC gate supply voltage VG+ and the off state value of drive signals J1 and J2 is the negative DC gate supply voltage VG−.
After the stepwise change of the input signal A followed by a high-to-low propagation delay tHLpd the drive signals J1 and J2 start to switch their voltage values. The first drive signal J1 starts decreasing its value from the on state voltage. The first power switch PT1 remains within the ohmic region until the drive signal J1 reaches the load current threshold voltage Vth@IL. At this point the first power switch PT1 comes into the active region and starts to switch off. The drain source voltage of the first power switch PT1 increases very fast to the DC link voltage being defined by the summation of power source voltages POWER+ and POWER−. This voltage change causes a current through the drain gate capacity CGD of the first power switch PT1 which holds the first drive signal J1 value constant until the time when the drain source voltage of the first power switch PT1 increases to the value of the DC link voltage. This effect is known as the so called Miller effect and appears simultaneously also on the second drive signal J2, as the drain source voltage change occurs also on the second power switch PT2. The second power switch PT2 is meanwhile still deeply in the cutoff region and there are no effects on the state of the second power switch PT2. After the voltage transition on the power switches PT1 and PT2 has been finished the first drive voltage J1 continues to decrease. Simultaneously with the decrease of the first drive voltage J1 the current through the first power switch PT1 starts decreasing and drops to a value 0 A at the time when the first drive voltage J1 becomes equal to the threshold voltage Vth. The first switch PT1 is off. The first drive voltage J1 keeps decreasing without any further influence to the state of the first power switch PT1. Since the second drive voltage J2 is still in the cutoff region and remains this way until it reaches the threshold voltage Vth both power switches PT1 and PT2 are off simultaneously for the interlock delay ID time.
Since the interlock delay ID time is generated automatically by mirroring of the drive signals J1 and J2 it is extremely short and stable. The drive signals J1 and J2 are mirrored with respect to the mirroring voltage value Vm at least for the transition time tt. The transition time tt represents the time interval in which the value of both drive signals J1 and J2 is lower than the threshold voltage Vth of power switches PT1 and PT2, which is ensured by the mirroring voltage Vm being lower than the threshold voltage Vth of the power switches PT1 and PT2. Therefore the mirroring voltage Vmis within the so called cutoff region. Within the cutoff region the power switches PT1 and PT2 are turned off.
The slope of the drive signals J1 and J2 and the mirror voltage Vm define the value of the interlock delay ID. The greater the slope is the shorter is the interlock delay ID and vice versa. The value of the interlock delay ID can reach down to 4 nsec while still remaining stable. The most important parameter for the interlock delay ID adjustment is the mirror voltage Vm. The value of the mirroring voltage Vm is adjustable as it is defined as the middle voltage between the DC gate supply voltages VG+ and VG− that can be controlled via the gate supply switching circuit 10 shown in FIG. 8. This short interlock delay ID prevents any cross-conduction through the power switches PT1 and PT2.
During the time when the second drive voltage J2 increases its value from the threshold voltage Vth to the load current threshold voltage Vth@IL the load current IL from the body diode DB is transferred to the FET cell FC of the second power switch PT2. The second power switch PT2 is turned on. The further increase of the second drive voltage J2 pushes the second power switch PT2 even deeper into the ohmic region.
The shaping of the ramp of the drive signals J1 and J2 is performed by the ramp generating circuit 62 and 62 formed by inductances L1 and L2 as shown in FIG. 4. In the switching time ts a part of the driving energy is stored in inductances L1 and L2. The switching time ts is defined as the time between the individual drive signal J1 or J2 changing its value from the on state voltage to the negative DC gate supply voltage VG− or vice versa from the off state voltage to the positive DC gate supply voltage VG+. The stored energy extends the slope of the first drive signal J1 under the negative DC gate supply voltage VG− and the slope of the second drive signal J2 over the positive DC gate supply voltage VG+ by causing an oversway. The over sway magnitude VOS+ and VOS− is in the embodiment shown in FIG. 4 defined by a damping resistors R1 and R2. After the oversway the second drive voltage J2 stays stable at the level of positive DC gate supply voltage VG+ and the first drive voltage J1 stays stable at the level of the negative DC gate supply voltage VG− and consequently the second power switch PT2 stays switched on whereas the first power switch PT1 stays switched off.
In the case of the switching shown as the right waveform in the FIG. 2 the drive signals J1 and J2 respond to the rising edge of the input signal A with a low-to-high propagation delay tLHpd. The second drive signal J2 starts decreasing its value from the on state voltage. In the voltage range from load current threshold voltage Vth@IL to threshold voltage Vth the body diode of the second power switch PT2 overtakes the load current IL from the FET-cell of the power switch PT2. The situation of the conducting states of the power switches PT1 and PT2 stays unchanged for the interlock delay ID time until the rising first drive signal J1 reaches the threshold voltage Vth.
The value of the interlock delay ID is the same as for the soft switching shown as the left side waveforms in FIG. 2, since the mirroring voltage Vm and the slope defined by inductances L1 and L2 of the drive signals J1 and J2 stay unchanged. Despite the electrical current flowing through the body diode DB of the second power switch PT2 the short value of the interlock delay ID prevents the charge carriers to occur. Therefore the body diode DB operates as an ideal diode without any reverse recovery effect for the time of the power switch PT1 overtaking the load current IL. The load current IL overtaking is finished when the first drive signal J1 reaches the load current threshold voltage Vth@IL.
The Miller effect occurs. The values of the drive signals J1 and J2 remain unchanged until the drain source voltage at the second power switch PT2 is increased to the DC link voltage and the drain source voltage at the first power switch PT1 falls to IL*Rds_onPT1. The on resistance Rds_onPT1 of the first power switch PT1 is the drain source resistance of the switched on power switch PT1. The further increase of the first drive voltage J1 pushes the first power switch PT1 even deeper into the ohmic region. The energy stored within the switching time ts prolongs the slope of the second drive signal J2 under the DC gate supply voltage VG− and the slope of the first drive signal J1 over the DC gate supply voltage VG+ by causing an over sway. After the signal over sway the first drive voltage J1 stays stable at the value of the positive DC gate supply voltage VG+ and the second drive voltage J2 stays stable at the value of the negative DC gate supply voltage VG− and consequently the first power switch PT1 stays switched on and the second power switch PT2 stays switched off.
The possible difference in propagation delays tHLpd and tLHpd according to the present invention has absolutely no influence on the interlock delay ID value. It only influences the value of the PWM ratio of the output PWM signal K with respect to the input PWM signal A.
FIG. 3 shows a flow chart of a possible embodiment of the method for driving a half bridge HB in terms of having a near zero interlock delay NZID time between an on state of the power switches PT1 and PT2 according to the present invention.
In a first step S1 the triggering signals F1 and F2 are generated. A first triggering signal F1 changes as a response to every odd step change of the input signal A its value in a stepwise from an on-voltage state to a positive DC gate supply voltage VG+ simultaneously with a second triggering signal F2 which changes as a response to every even step change of the input signal its value in a stepwise from an off-voltage state to a negative DC gate supply voltage (VG−).
In the second step S2a shaping of the triggering signals F1 and F2 into the drive signals J1 and J2 is performed. The drive signals J1 and J2 switch the corresponding power switches PT1 and PT2 between the on and off states. The drive signals J1 and J2 are mirrored with respect to one another along the mirroring voltage Vm at least within the transition time tt. The mirroring voltage value Vm is the voltage value applied to the control electrode of the power switches PT1 and PT2 that ensures both power switches PT1 and PT2 to be within the cutoff region. The transition time tt is defined as the time when none of the power switches PT1,PT2 is in the so-called ohmic region. A power switch is in the ohmic region, when it is driven by a drive signal (J1 or J2) applied to the control electrode being larger as needed for ensuring the conduction of the load current IL. In the case of BJT transistors this region is called saturation.
A possible embodiment of the driving circuit 1 according to the present invention for driving a Si MOSFET is illustrated in FIG. 4. The driving circuit 1 of FIG. 4 comprises a differentiating circuit 4, a magnetic triggering circuit T1, triggering circuits 51 and 52, ramp generating circuits 61 and 62, a magnetic coupling circuit T3 and resistors R1 and R2.
The input signal A of the driving circuit 1 is connected to the differentiating circuit 4, which generates pulsed signals B and C. The waveforms of the pulsed signals B and C are shown as a signal diagram in the second and third time curve of FIG. 5. A pulse of the pulsed signal B is generated in response to the rising edge of the input signal A and a pulse of the pulsed signal C is generated in response to falling edge of the input signal A. The duration of these pulses is equal to an establishment time te, which is either predetermined or defined simultaneously with the operation according to a power switch state information IT3. In order to prevent a voltage drift of the drive signals J1 and J2 the differentiating circuit 4 can generate maintenance pulses with the same polarity as the last pulse before. Maintenance pulses hold the drive signals J1 and J2 stable for an infinite time, especially in the case of a 0% or 100% duty cycle. A maintenance pulse is shown on the pulsed signal B in the second time curve of FIG. 5. Each pulse of the pulsed signals B and C is amplified within the differentiating circuit 4 to a value, which has to be high enough for the pulsed signals B and C to control the triggering switches Q1 and Q2 through the magnetic triggering circuit T1 without any additional amplification.
The magnetic triggering circuit T1 comprises a transformer with one primary and four secondary windings. Each side of the primary winding is connected to receive a pulsed signal B or C and each secondary winding is connected to one triggering transistor Q1 or Q2 of the triggering circuits 51 and 52. The magnetic triggering circuit T1 converts the pulsed signals B and C to a bipolar trigger signal BC with the magnitude VTr1 and VTr2 being higher than the bipolar threshold voltage VthQ1 and VthQ2 of a corresponding triggering switch Q1 or Q2. The quality of the magnetic triggering circuit T1 ensures a demagnetization over sway VOS1 and VOS2 value being lower than the bipolar threshold voltage VthQ1 and VthQ2. The theoretical waveform of the bipolar trigger signal BC is shown as the fourth time curve of FIG. 5 and the real waveform of the bipolar trigger signal BC is shown as the fifth time curve in FIG. 5. The winding orientation of the magnetic triggering circuit T1 is provided in such a way to ensure that the active state of the pulsed signal B does turn on the first triggering transistor Q1 of the first triggering circuit 51 and the second triggering transistor Q2 of the second triggering circuit 52 simultaneously and to ensure that the active state of the pulsed signal C does turn on the second triggering transistor Q2 of the first triggering circuit 51 and the first triggering transistor Q1 of the second triggering circuit 52 simultaneously.
The triggering switches Q1 and Q2 can be any electrically controlled switching element or a combination of such electrically controlled switching elements. The trigger switches Q1 and Q2 are serial connected and powered via the DC gate supply circuit. The first trigger switches Q1 connect the common point of the trigger switches Q1 and Q2 to the positive DC gate supply voltage VG+. The second trigger switches Q2 connect the common point of the trigger switches Q1 and Q2 to the negative DC gate supply voltage VG−. The voltage at the common point of the triggering switches Q1 and Q2 of first triggering circuit 51 is the first triggering signal F1. The voltage at the common point of the triggering switches Q1 and Q2 of the second triggering circuit 52 forms the second triggering signal F2. The first triggering signal F1 is shown as the sixth time curve in FIG. 5 and the second triggering signal F2 is shown as the seventh time curve in FIG. 5. The positive DC gate supply voltage VG+ represents the on-state voltage of the trigger signals F1 and F2 and the negative DC gate supply voltage VG− represents the off-state voltage of the trigger signals F1 and F2. Both triggering signals F1 and F2 are grounded to a DC gate supply voltage VGND by means of a security resistance Roff. The simultaneity of the switching of the first triggering switch Q1 of the first triggering circuit 51 and the second triggering switch Q2 of second triggering circuit 52 guaranteed by the magnetic triggering circuit T1 ensures a complementary switch-on of the first triggering signal F1 and switch-off of the second triggering signal F2. The simultaneity of the switching of the second triggering switch Q2 of the first triggering circuit 51 and the first triggering switch Q1 of the second triggering circuit 52 guaranteed by the magnetic triggering circuit T1 ensures a complementary switch-on of the second triggering signal F2 and switch-off of the first triggering signal F1.
The triggering circuits 51 and 52 both include a DC gate supply circuit 7. The DC gate supply circuit 7 supplies DC gate supply voltages VG+, VG− and VGND to the triggering switches Q1 and Q2. A first energy E1 powers the DC supply circuit 7 of the first triggering circuit 51 and a second energy E2 powers the DC supply circuit 7 of the second triggering circuit 52.
The security resistance Roff ensures that the voltage of the control electrode G is 0V, which in turn ensures that all power switches PT1 and PT2 of the normally off type are switched off and that the turn off of the power switches PT1, PT2 is executed in the absence of the bipolar pulsed signal BC. The security resistance Roff causes a voltage drop of the triggering signals F1 and F2 in the time of an inactive bipolar pulsed signal BC. The maintenance pulse returns the dropped voltage value of the triggering signals F1 and F2 back to the previous value.
Conversion of the first triggering signal F1 into the first drive signal J1 is performed by a corresponding first ramp generating circuit 61, which is formed by an energy buffer component comprising an inductor L1. Conversion of the second triggering signal F2 into the second drive signal J2 is performed by a corresponding second ramp generating circuit 62, which is formed by an energy buffer component comprising an inductor L2. The first drive signal J1 is applied to the control electrode G of the first power switch PT1 and the second drive signal J2 is applied to the control electrode G of the second power switch PT2. The above mentioned conversion of the triggering signals F1 and F2 into corresponding drive signal J1 and J2 causes a difference in the integral of the triggering signal F1 or F2 and the integral of the corresponding drive signal J1 or J2. The first drive signal J1 and the voltage difference area AVD are shown as the sixth time curve of FIG. 6. The stepwise change of the triggering signal F1 or F2 causes a voltage difference on the corresponding inductor L1 or L2 with a maximum value of the voltage difference of the stepwise change of the triggering signal F1 or F2, which drops to 0V within the switching time ts. Therefore a current IL1, IL2 through the corresponding inductor L1, L2 starts increasing and after that the switching time ts reaches a maximum value. The maximum values of currents IL1 and IL2 are defined by the voltage difference area AVD and the corresponding inductor L1 or L2. The inductor current IL1 is shown as the fifth curve of FIG. 6.
The energy stored in the inductors L1 and L2 due to the corresponding current IL1 and IL2 generates an over sway on the corresponding drive signals J1 or J2. A first resistor R1 is connected in parallel to the first ramp generating circuit 61 and a second resistor R2 is connected in parallel to the second ramp generating circuit 62 in order to prevent the rising of the corresponding drive signal J1 and J2 after the over sway. The resistor current IR1 is shown as the fourth curve of FIG. 6.
In order to ensure a precise mirroring of the drive signals J1 and J2 the mirroring of the voltage difference between the triggering voltage F1 and the first drive voltage J1 with respect to the voltage difference between the triggering voltage F2 and the second drive voltage J2 is performed by means of a magnetic coupling circuit T3. In the possible embodiment of FIG. 4 the magnetic coupling circuit T3 is formed by one transformer having its primary winding connected in parallel to the first ramp generating circuit 61 and having its secondary winding connected in parallel to the second ramp generating circuit 62. The primary to secondary winding ratio is −1. The mirroring is performed by an equalizing current Ieq flowing through the magnetic coupling circuit T3 windings during the time when one of the triggering switches Q1 or Q2 of both triggering circuits 51 and 52 conduct in any direction. Equal values of the inductance of the inductors L1 and L2 as well as equal values of the resistance of the resistors R1 and R2 reduce the equalizing current Ieq through the magnetic coupling circuit T3.
Through an additional winding of the magnetic coupling circuit T3 the establishment status information Infe is transferred to the differentiating circuit 4. The additionally winding represents the voltage on the ramp generating circuits 61 and 62, which forms simultaneously the establishment status information Infe. Thus, the establishment status information is an indicator of the current switch state of the corresponding power switch PT1 or PT2. A significant information about the switching status is the time tIe in which the establishment status information Infe reaches a value 0 after having reached its maximum value. By having the establishment time te larger than the time tIe the driving energy EL1 or EL2 needed for the switching execution is higher than in the case of having the establishment time te shorter than the time tIe. The over sway value after time tIe represents the oversway in the drive signals J1 and J2. By shortening the establishment time te the over sway can be hold in proper ranges without provision of corresponding resistors R1 and R2. FIG. 7 shows a comparison of a large establishment time te on the left and a short establishment time te on the right side. The first time curve in FIG. 7 shows a bipolar pulsed signal BC with a marked up establishment time te. The second time curve in FIG. 7 shows the first triggering signal F1. The third time curve in FIG. 7 shows an establishment time information Infe with a marked up time tIe. The fourth time curve in FIG. 7 shows the current IL1 with a marked up driving energy EL1. The fifth time curve in FIG. 7 shows the first drive signal J1.
The energy for driving the power switches PT1 and PT2 is supplied via the powering circuit 13 as shown in FIG. 8. The powering circuit 13 comprises at least a supply switching circuit 10 and a power supply magnetic circuit T2. The gate switching circuit 10 generates a signal O by switching a gate power source voltage N by means of a switch SW1 in a ratio defined by a gate power ratio control signal M. The coupling capacitor C5 prevents the power supply magnetic circuit T2 to saturate. The signal O carries an energy ET2, which is transferred through the magnetic coupling circuit T2 to the DC gate supply circuits 7. The amplitude of the signal O is defined by the magnitude of the gate power source voltage N. The PWM ratio of the signal O is defined by the duty cycle of the gate power ratio control signal M. A bigger duty cycle of the gate power ratio control signal M simultaneously lowers the positive DC gate supply voltage VG+ and increases the negative DC gate supply voltage VG− and vice versa a lower duty cycle of the gate power ratio control signal M simultaneously increases the positive DC gate supply voltage VG+ and reduces the negative DC gate supply voltage VG−. This is ensured by the power supply magnetic circuit T2 due to its property of equalizing the positive voltage integral areas Φ1 and Φ2 and the negative voltage integral areas Φ1′ and Φ2′ of the signal O. The amplitude and the PWM ratio of signal O defines the value of the DC gate supply voltages VG+ and VG−. The values of the DC gate supply voltages VG+ and VG− have both a linear dependency of the gate power source voltage N. Apparently, by controlling the values of the gate power ratio signal M and the gate power source voltage N the DC gate supply voltages VG+ and VG− can be adapted.
FIG. 9 shows the time diagrams of two examples of setting the DC gate supply voltages VG+ and VG−. Both examples are shown separately with a dotted line. On the left side of the dotted line an example of a bigger duty cycle of the gate power ratio signal M and a higher value of the gate power source voltage N is shown. On the right side of the dotted line an example of a smaller duty cycle of the gate power ratio signal M and a lower value of the gate power source voltage N is shown. The first time curve of the FIG. 9 shows the gate power ratio signal M, the second time curve of the FIG. 9 shows gate power source voltage N and the third time curve of the FIG. 9 shows the signal O with illustrated DC gate supply voltages VG+ and VG− as a response of the corresponding gate power ratio signal M and the corresponding gate power source voltage N.
A detailed schematic diagram of the DC gate supply circuit 7 is illustrated in FIG. 8. Each DC gate supply voltage VG1+, VG+, VG−, VGND is generated by one half wave rectifier D1 or D2 or D3 and a corresponding smoothing capacitor C1 or C2 or C3. The DC gate supply voltages VG+ and VG− are powered though the same winding. In order to generate the DC gate supply voltage VG+ an additional tap T′ on the corresponding secondary winding of the power supply magnetic circuit T2 is used.
The signal O voltage output by the secondary winding of the power supply magnetic circuit T2 is used to define a shut down information InfSD. In the possible embodiment as illustrated in the fourth time curve in FIG. 9a signal is generated through the gate power ratio control signal M causing a change of the frequency with the same PWM ratio at least for one period of the signal O. An operating period time top reduces to a shut down period time tsd, which is detected by the shut down circuit 8. The connection of the shut down circuit 8 is shown in FIGS. 15-16.
By setting the values of the gate power ratio control signal M and the gate power source voltage N the DC gate source voltages VG+ and VG− and consequently also the average voltage value of the DC gate source voltages VG+ and VG−, namely the mirroring voltage Vm is defined. Setting of the DC gate source voltages VG+ and VG− and the mirroring voltage Vm is limited by the condition that the mirroring voltage Vm needs to be below the threshold voltage Vth value. The greater the difference between the mirroring voltage Vm and the threshold voltage Vth is the greater is the interlock delay ID value and vice versa, the smaller the difference between the mirroring voltage Vm and the threshold voltage Vth is the smaller is the interlock delay ID value. The time curves of the drive signals J1 and J2 in FIG. 10 of one possible embodiment show the difference in the interlock delay ID for two different values of the DC gate supply voltages VG+ and VG−.
FIG. 11 shows a possible embodiment of a circuit which ensures a stable interlock delay ID regardless the temperature changes of the power switches PT1 and PT2. With the temperature increase a drop in the threshold voltage Vth occurs as shown in the first time curve of FIG. 12. Therefore the mirroring voltage Vm needs to change with the same voltage difference over the entire temperature range of the power switches PT1 and PT2.
In the possible embodiment shown in FIG. 11 a temperature influence adapting circuit 11 and a DC/DC converter is added to the powering circuit 13. In order to achieve a constant voltage difference Vdif despite the variations in a junction temperature Tj of the power switches PT1 and PT2 the temperature influence adapting circuit 11 adjusts the gate power ratio control signal M and controls a DC/DC converter 12 so that the DC/DC converter 12 generates a gate power source voltage N with respect to a temperature information InfT. The voltage difference Vdif is the voltage difference between the mirroring voltage Vm and the threshold voltage Vth. The temperature information InfT includes an information about the junction temperature Tj of the power switches PT1 and PT2.
The first time curve in FIG. 12 illustrates a possible example of the threshold voltage Vth dependence on the junction temperature Tj. The second and third time curves in FIG. 12 show the gate power ratio control signal M and the gate power source voltage N as a response to the temperature information InfT with respect to the junction temperature Tj. The fourth time curve in FIG. 12 shows the voltage difference Vdif, which is constant throughout the entire junction temperature Tj range of the power switches PT1 and PT2 and the corresponding DC gate supply voltages VG+ and VG− that ensure such a constant voltage difference Vdif.
In the possible embodiment of the FIGS. 13a-13d an energy return for on switching of the first power switch PT1 is demonstrated and in FIGS. 14a-14d an energy return for off switching of the first power switch PT1 is demonstrated. Only the elements that are included in the energy transfer of the first power switch PT1 are shown. The circuit in FIGS. 14a-14d has an additional tap T and additional capping diodes D6 and D7 added with respect to the part of the driving circuit 1 that controls the first power switch PT1 in the embodiment shown in FIG. 4. The magnetic coupling circuit T3 has an additional tap T on the winding connected to inductor L1 of the first ramp generating circuit 61 to form an autotransformer function. The position of the tap T is set closer to the side of the winding, which is connected to the control electrode G of the first power switch PT1 and forms a coupling ratio RC<1 between 0.5 and 1. The capping diode D6 is connected to the tap T to receive the DC gate supply voltage VG+ of the DC gate supply circuit 7. The capping diode D7 is connected between the negative DC gate supply voltage VG− of the DC gate supply circuit 7 and the tap T.
In the possible embodiment shown in FIG. 13a in the establishment time te duration the triggering switch Q1 is switched on and a gate current IG1 transfers the driving energy from the smoothing capacitor C1 to capacitor CGS. As the gate current IG1 flows via the inductor L1 the value of the energy stored in the inductor L1 is equal to the value of the energy stored in the capacitor CGS. After the triggering switch Q1 has switched off as shown in FIG. 13b the gate current IG1 still flows due to the energy stored in the inductor L1 within the establishment time te. The energy is transferred from inductor L1 to the capacitor CGS and the smoothing capacitor C2. The part of the energy that is returned to the smoothing capacitor C2 is the energy, which is returned back to the DC power supply circuit 7. With the drive signal J1 voltage value reaching the DC gate supply voltage VG+ divided by the coupling ratio RC<1 the gate current IG1 drops to 0 and the energy, which is still stored in the inductor L1 is transferred through the capping diode D6 to the smoothing capacitors C1 and C2 of the DC power supply circuit 7 as shown in FIG. 13c. After the entire energy in the inductor L1 has been transferred the energy stored in capacitor CGS due to the oversway is returned back to the smoothing capacitor C1 of the DC gate supply circuit 7 through the capping diode D6 as shown in FIG. 13d. 
In the possible embodiment shown in the FIG. 14a in the establishment time te duration the triggering switch Q2 is switched on and a gate current IG1 transfers the driving energy from smoothing capacitor C2 to capacitor CGS. As the gate current IG1 flows via the inductor L1 the value of the energy stored in the inductor L1 is equal to the value of the energy stored in the capacitor CGS. After the triggering switch Q2 has switched off as shown in FIG. 14b the gate current IG1 still flows due to the energy stored in the inductor L1 within the establishment time te. The energy is transferred from inductor L1 to capacitor CGS and the smoothing capacitor C1. The part of the energy that is returned to the smoothing capacitor C1 is the energy, which is returned back to the DC power supply circuit 7. With the drive signal J1 voltage value reaching the DC gate supply voltage VG+ divided by the coupling ratio RC<1 the gate current IG1 drops to 0 and the energy, which is still stored in the inductor L1 is transferred through the capping diode D7 to the smoothing capacitors C1 and C2 of the DC power supply circuit 7 as shown in FIG. 14c. After the entire energy in the inductor L1 has been transferred the energy stored in capacitor CGS due to the oversway is returned back to the smoothing capacitor C2 of the DC gate supply circuit 7 through the capping diode D7 as shown in FIG. 14d. 
In a possible embodiment the power switches PT1 and PT2 are formed by switching elements such as BJTs or JFETs, which demand a constant current into the control electrode G to hold the power switch PT1 and PT2 on or off.
FIG. 15 shows a possible embodiment of the driving circuit 1 driving the half bridge HB connected BJTs as power switches PT1 and PT2. FIG. 16 shows a possible embodiment of the driving circuit 1 driving the in half bridge HB connected JFETs as power switches PT1 and PT2. In the possible embodiments of the FIGS. 15 and 16 a bistable circuit 9 is connected to the triggering circuits 51 and 52 in order to provide a constant current flowing to the control electrode G of the power switches PT1 and PT2. The bistable circuit 9 is powered via the DC gate supply circuit 7 with the DC gate supply voltages VG1+ and VG− and consists of two current generators Ion and Toff, a comparator X and a bistable circuit resistor Rbc. The comparator X has a hysteresis of approximately 1V with respect to DC gate supply voltage VGND. At the time the drive signals J1 and J2 being within the hysteresis area the comparator X does not switch the current generators Ion or Toff on. At the time when the drive voltage is over the hysteresis the comparator X switches the current generator Ion on and at the time when the drive voltage is below the hysteresis the comparator X switches the current generator Ioff on. A bistable circuit resistor Rbc is added in the feedback loop of the comparator X to prevent the current reduction at the time when the shut down circuit is turned on. The current generator Ion is powered via the DC gate supply circuit 7 with the DC gate supply voltage VG1+ and generates a preset current value needed to hold the on-state of the corresponding power switch PT1 and PT2. The position of the tap T′ on the secondary winding of the power supply magnetic circuit T2 defines the minimum value of the DC gate supply voltage VG1+ that still holds the voltage drop on the current generator Ion high enough for the current generator Ion to be able to generate the preset current in all working condition. A recommended voltage drop on the current generator Ion is up to 2 V. The current generator Ioff is powered via the DC gate supply circuit 7 with the negative DC gate supply voltage VG− and generates a current needed to hold the off-state of the corresponding power switch PT1 and PT2. The current generated by the current generator Ioff is predefined and has a value within the mA region and can therefore be replaced with a resistor.
For a fast turning off of the power switches PT1 and PT2 a shut down circuit 8 is added to the triggering circuits 51 and 52. The shut down circuit 8 includes a decoding circuit, which detects a change in the frequency of the signal O and as a response to a detected shut down period time tsd deactivates the bistable circuit 9, connects the corresponding drive signal J1 or J2 to the DC gate supply voltage VGND and consequently turns the corresponding power switch PT1 or PT2 off.
In a possible embodiment in FIG. 15 the driving circuit 1 controls BJTs as power switches PT1 and PT2, therefore the on state voltage VPT1on and VPT2on is lower than the DC gate supply voltage VG+ due to the properties of the BJTs. At the time the triggering switch Q2 switches on a voltage difference between the DC gate supply voltage VG− and on state voltage Von occurs on the corresponding ramp generating circuit 61 or 62. At the time the triggering switch Q1 switches on a voltage difference between the DC gate supply voltage VG+ and the off state voltage Von occurs at the corresponding ramp generating circuits 61 or 62. The short circuit current through the magnetic coupling circuit T3 due to the voltage difference that thus occurs between ramp generating circuit 61 or 62 is prevented by a blocking diode D9 added to the triggering circuits 51 and 52 between the triggering switch Q2 and the negative DC gate supply voltage VG− of the DC gate supply circuit 7. Only an energy return through the capping diode D7 is possible. Therefore only the capping diode D7 is used for the energy return to the DC power supply circuit 7. The part of energy stored in the inductors L1 and L2 during the establishment time te is only returned to the DC gate supply circuit 7, which belongs to the second power switch PT1 or PT2 that is turning off. Therefore the driving energy EL1 is during the switch off of the power switch PT2 transferred through the magnetic coupling circuit T3 and the driving energy EL2 is during the switch off of the first power switch PT1 transferred through the magnetic coupling circuit T3.
The FIGS. 17 and 18 show a response of the characteristic time curves with respect to the input signal A shown in the first time curve of the FIG. 17. The time markers a-l are equal for FIG. 17 and FIG. 18. The time markers a and g present the beginning of the switching. The time markers b and h present the time when the bistable circuit 9 turns off the current generator Ion. The time markers c and i represent the time when the bistable circuit 9 turns on the current generator Ion. The time markers d and j represent the time when the triggering switch Q2 stops conducting due to the drive signals J1 and J2 being equal to the negative DC gate supply voltage VG−. The time markers e and k represent the time after the establishment time te has expired. The time markers f and l represent the time when the energy return process is finished.
The second time curve of the FIG. 17 shows a signal diagram of the bipolar pulsed signal BC with a marked up establishment time te. The third time curve of the FIG. 17 shows a signal diagram of the current IQ1_1 of the triggering switch Q1 of the first triggering circuit 51. The fourth time curve of the FIG. 17 shows a signal diagram of the current IQ2_1 of the triggering switch Q2 of the second triggering circuit 51, which is equal to the current ID9_1 of the blocking diode D9 of the same triggering circuit 51. The fifth time curve of the FIG. 17 shows a signal diagram of the current ID7_1 of the capping diode D7 of the first triggering circuit 51. The sixth time curve of the FIG. 17 shows signal diagram of the first triggering signal F1. The seventh time curve of the FIG. 17 shows signal diagram of the current IG1 flowing to the control electrode of the first power switch PT1 with a marked up current ION1 of the current generator Ion. The eighth time curve of the FIG. 17 shows the signal diagram of the first drive signal J1.
The first time curve of the FIG. 18 shows the signal diagram of the establishment time information Infe. The second time curve of the FIG. 18 shows the signal diagram of the current IQ1_2 of the triggering switch Q1 of the second triggering circuit 52. The third time curve of the FIG. 18 shows the signal diagram of the current IQ2_2 of the triggering switch Q2 of the second triggering circuit 52, which is equal to the current ID9_2 of the blocking diode D9 of the same triggering circuit 52. The fourth time curve of the FIG. 18 shows signal diagram of the current ID7_2 of the capping diode D7 of the second triggering circuit 52. The fifth time curve of the FIG. 17 shows signal diagram of the second triggering signal F2. The sixth time curve of the FIG. 18 shows signal diagram of the current IG2 flowing to the control electrode of the second power switch PT2 with a marked up current ION1 of the current generator Ion. The eighth time curve of the FIG. 17 shows the signal diagram of the drive signals J1 and J2 with a marked up interlock delay ID. The dot hatched area on the FIG. 18 shows the on and off states of the power switches PT1 and PT2.
In a possible embodiment in FIG. 19 the driving circuit 1 controls normally on JFETs as power switches PT1 and PT2. In order to prevent the cross conduction through the power switches PT1 and PT2 in the time before and during the establishment of the operation conditions of the driving circuit 1 the cascode power switch PTc is connected in a cascode with the first power switch PT1 and a cascode power switch PTc connected with the second power switch PT2 is added. A cascode power switch PTc is formed by a low voltage and low on resistance MOSFET. The voltage of the cascode power switch PTc needs to be high enough to hold the power switch PT1 or PT2 in the cutoff region when the voltage on the control electrode G of the power switch PT1 or PT2 reaches a positive DC gate supply voltage VG+. The on resistance of the cascode power switch PTc should be in a mΩ range in order for the cascode power switch PTc not to essentially contribute to the conductive losses of the half bridge HB. In the possible embodiment of FIG. 19 an off current generating resistor RIG-substitutes the off current generator Ioff and a cascode switching circuit 14 is added to the triggering circuits 52 and 52 in order to control the cascode power switch PTc. The cascode switching circuit 14 ensures an off state of the cascode power switch PTc during the establishment of the operation conditions of the driving circuit 1 by holding the control electrode G of the cascode power switch PTc connected to the source S of the cascode power switch PTc. In the time when the DC gate supply circuit 7 has not yet established the final values of the DC gate supply voltages VG+, VG1+ and VG− the control electrode G of the power switch PT1 and PT2 is connected through the off current generating resistor RIG− to the DC gate supply voltage VG− and through the half wave rectifier D2 and the corresponding secondary winding of the power supply magnetic circuit T2 back to the source S of the cascode power switch PTc. Thus, the current through the corresponding power switch PT1 or PT2 is blocked. When the negative DC gate supply voltage VG− has reached its nominal value the cascode switching circuit 14 turns the cascode power switch PTcon. The corresponding power switch PT1 or PT2 remains in the cutoff region despite the cascode power switch PTc being turned on as the voltage at the control electrode G of the corresponding power switch PT1 or PT2 remains lower than the threshold voltage Vth, namely at the level of the negative DC gate supply voltage VG−.
In the possible embodiment of FIG. 20 SiC FETs are provided as power switches PT1 and PT2. SiC FETs are limited with a negative voltage to a maximum negative driving voltage −10V, have the threshold voltage Vth in the range up to 3V and need up to +20V to be completely in the ohmic region. A blocking diode D8 and the only the capping diode D6 is added to the triggering circuit 51 and 52. The DC gate supply voltage VG− is defined to be close to the maximum negative driving voltage, for example −8V for a −10V device limit. The mirroring voltage Vm is defined to be within the cutoff region of the power switches PT1 and PT2. The positive DC gate supply voltage VG+ is equal Vm+(Vm−VG−), where Vm presents the mirroring voltage and VG− presents the DC gate supply voltage VG−. On state voltage Von is defined to be close to the maximum positive driving voltage, for example 20V for a −22V device limit with the autotransformer coupling ratio RC<1 through the equation (VG+−VG−)/(Von−VG−), where Von presents the on state voltage, VG+ represents the positive DC gate supply voltage and VG− represents the negative DC gate supply voltage.
The driving energy EL1 and EL2 is in the possible embodiment of the FIG. 20 returned to the DC power supply circuit 7 of the triggering circuit 51 or 52 at the time of the corresponding power switch PT1, PT2 is switching on. The driving energy EL1 and EL2 is transferred through the magnetic coupling circuit T3 via the capping diode D6 of the first triggering circuit 51 connected to the corresponding tap T of the magnetic coupling circuit T3 and returned back to the DC gate supply circuit 7 of the first triggering circuit 51 at the time when the power switch PT1 is in the switch on transition. The driving energy EL1 and EL2 is transferred through the magnetic coupling circuit T3 via the capping diode D6 of the second triggering circuit 52 connected to the corresponding tap T of the magnetic coupling circuit T3 and returned back to the DC gate supply circuit 7 of the second triggering circuit 52 at the time when the power switch PT2 is in the switch on transition.
FIG. 21 illustrates simplified time curves of a switch off transition of the first power switch PT1 and a switch on transition of the second power switch PT2. At time marking t0 the triggering switch Q1 of the first triggering circuit 51 and the triggering switch Q2 of the second triggering circuit 52 is switched on. The first drive signal J1 starts to drop. The second drive signal J2 remains stable on the value of the off state voltage Voff due to the blocking diode D8. At the time marking t1 the voltage on inductor L1 drops to a value |VG−|+VG+, where VG− presents the negative DC gate supply voltage and the VG+ presents the positive DC gate supply voltage. The first drive signal J1 starts to increase. At the time marking t2 the first drive voltage J1 reaches the load current threshold voltage Vth@IL and the output voltage K starts to drop. The Miller effect becomes effective. The dropping in the output voltage K after time marking t3 speeds up due to the reduction of the gate-drain capacity of the first power switch PT1 when the gate drain voltage of the first power switch PT1 overtakes the gate source voltage of the power switch PT1. The difference between the gate drain capacities of the power switches PT1 and PT2 causes an equalizing current Ieq flowing through the magnetic coupling circuit T3 between the time markings t2 and t3. The load current ILPT1 transfers from the FET cell FC of the first power switch PT1 to the body diode DB of the second power switch PT2 during the time when the first drive signal J1 drops to the threshold voltage Vth which is reached at the time marking t5. The first power switch PT1 is off. At the time marking t6 the second drive signal J2 increases to the threshold voltage Vth. The load current ILPT2 is transferred from the body diode DB to the FET cell FC of the second power switch PT2 between the time markings t6 and t7. The energy stored in the inductors L1 and L2 cause the second drive voltage J2 to increase to on state voltage Von reached at the time marking t8. The energy stored in the inductor L2 is transferred to the gate source capacity of the second power switch PT2 through the magnetic coupling circuit T3. This energy transfer is shown by the equalizing current Ieq flowing between the time markings t7 and t8.
The first time curve of FIG. 21 shows the signal diagram of the drive signals J1 and J2. The second time curve of the FIG. 21 shows the signal diagram of the equalizing current Ieq. The third time curve of FIG. 21 shows the signal diagram of the output signal K. The fourth time curve of FIG. 21 shows the signal diagram of the load current ILPT1 flowing through the first power switch PT1. The fifth time curve of the FIG. 21 shows the signal diagram of the current flowing through the body diode DB2 of the second power switch PT2. The sixth time curve of the FIG. 21 shows the signal diagram of the load current ILPT2 of the second power switch PT2.
In the possible embodiment of FIG. 22 the first power switch PT1 consists of serially connected switching elements PT1a, PT1b . . . PT1n and the second power switches PT2 consists of serially connected switching elements PT2a, PT2b . . . PT2n. The first triggering circuit 51 comprises triggering circuits 51a, 51b . . . 51n and the second triggering circuit 52 comprises triggering circuits 52a, 52b . . . 52n. The triggering circuits 51a, 51b . . . 51n, 52a, 52b . . . 52n are driven with the magnetic triggering circuit T1. The magnetic triggering circuit T1 comprises six transformers each having one primary and two secondary winding. The first ramp generating circuit 61 consists of 61a, 61b . . . 61n and the second ramp generating circuit 62 consists of 62a, 62b . . . 62n. Each triggering circuit 51a, 51b . . . 51n, 52a, 52b . . . 52n is connected to a corresponding ramp generating circuit 61a, 61b . . . 61n, 62a, 62b . . . 62n being connected to the corresponding switching element PT1a, PT1b . . . PT1n, PT2a, PT2b . . . PT2n and comprising the inductor L1. The resistor R1 is connected in parallel to every ramp generating circuit 61a, 61b . . . 61n, 62a, 62b . . . 62n. The magnetic coupling circuit T3 consists of six transformers each having one of its windings connected to one of the ramp generating circuits 61a, 61b . . . 61n, 62a, 62b . . . 62n and the remaining windings connected in parallel between each other. A coupling ratio between the windings of magnetic coupling circuit T3 connected to any pair of the ramp generating circuits 61a, 61b . . . 61n is 1 and a coupling ratio between windings of the magnetic coupling circuit T3 connected to any pair of the ramp generating circuits 62a, 62b . . . 62n is 1. The coupling ratio between the windings of the magnetic coupling circuit T3 being connected to the ramp generating circuits 61n and 62n is −1.
In a possible embodiment of the FIG. 23 the driving circuit 1 according to the present invention controls the first power switch PT1, which consists of two switching elements PT1a and PT1b and the second power switch PT2, which consists of two switching elements PT2a and PT2b. The switching elements PT1a and PT2a are also connected in a half bridge and the switching elements PT1b and PT2b are connected in a half bridge. The switching elements PT1a and PT2b are powered with the power supply POWER+ and switching elements PT2a and PT1b are powered with the power supply POWER+. The load is connected between a common point of the switching elements PT1a and PT2b and a common point of the switching elements PT2a and PT1b. The magnetic triggering circuit T1 has one transformer with one primary and eight secondary windings.
The first triggering circuit 51 consists of two triggering circuits 51a and 52b and the second triggering circuit 52 consists of two triggering circuits 52a and 51b. The first ramp generating circuit 61 consists of two ramp generating circuits 61a and 62b and the second ramp generating circuit 62 consists of two ramp generating circuits 62a and 61b. The triggering circuit 51a controls the switching element PT1a through the ramp generating circuit 61a, the triggering circuit 52a controls the switching element PT2a through the ramp generating circuit 62a, the triggering circuit 51b controls the switching element PT1b through the ramp generating circuit 61b and the triggering circuit 52b controls the switching element PT2b through the ramp generating circuit 62b. A resistor R1 is connected in parallel to the ramp generating circuits 61a and 62b. A resistor R2 is connected in parallel to the ramp generating circuits 62a and 61b. The magnetic coupling circuit T3 is formed by two transformers each having windings connected in parallel to the ramp generating circuits 61a, 62a, 61b and 62b as shown in FIG. 23.
ABBREVIATIONS


    NZID near zero interlock delay
    ID interlock delay
    DT dead time
    PT power switch
    PTc cascade power switch
    PT1, PT2 power switches
    PT1a, PT1b . . . PT1n switching elements of PT1
    PT2a, PT2b . . . PT2n switching elements of PT2
    HB half bridge
    DB body diode
    FC FET cell
    S1 first step
    S2 second step
    A input signal
    B, C pulsed signals
    BC bipolar pulsed signal
    F1, F2 triggering signals
    J1, J2 drive signals
    O signal
    N gate power source voltage
    M gate power ratio control signal
    B1, B2 signals
    P1, P2 signals
    B1′, B2′ signals
    1 driving circuit
    4 differentiating circuit
    51, 52 triggering circuits
    51a, 51b, . . . , 51n triggering circuits within 51 
    52a, 52b, . . . , 52n triggering circuits within 52 
    62, 62 ramp generating circuits
    61a, 61b, . . . , 61n triggering circuits within 61 
    62a, 62b, . . . , 62n triggering circuits within 62 
    7 DC gate supply circuit
    8 shut down circuit
    9 bistable circuit
    10 supply switching circuit
    11 temperature influence adapting circuit
    12 DC/DC converter
    13 powering circuit
    14 cascode switch control circuit
    T1 magnetic triggering circuit
    T2 power supply magnetic circuit
    T3 magnetic coupling circuit
    Del delay unit
    DR1, DR2 galvanically isolated driving circuits
    RC1, RC2 circuits
    DC′ power supply
    InfT temperature information
    Infn power switch state information
    InfSD shut down information
    Infe establishment time information
    Q1, Q2 triggering switches
    L1, L2 inductors
    R1, R2 damping resistors
    D6, D7 capping diodes
    D8, D9 blocking diodes
    CGD drain gate capacitor of power switch
    Rds_onPT1 resistor
    Roff security resistor
    Rbc bistable circuit resistor
    RIG− off current generating resistor
    X comparator
    D1, D2, D3 half wave rectifiers
    C1, C2, C3 smoothing capacitors
    SW1 switch
    C5 coupling capacitor
    RG1, RG2 resistor
    Dbs bootstrap diode
    I1, ILPT1, ILPT2 load current
    IL1, IL2 inductor current
    IR1, IR2 resistor current
    Ieq equalizing current
    VG+ DC gate supply voltage
    VG− DC gate supply voltage
    VGND DC gate supply voltage
    Vth@IL load current threshold value
    POWER+ power source voltage
    POWER− power source voltage
    Vth threshold voltage
    Vth@IL load current threshold voltage
    Vth@Tj junction temperature threshold voltage
    Vm mirroring voltage
    VOS+, VOS− over sway magnitudes
    VthQ1, VthQ2 bipolar threshold voltage
    VPT1on PT1 on state voltage
    VPT2on PT2 on state voltage
    VPT1off PT1 off state voltage
    VPT2off PT2 off state voltage
    Vdif voltage difference
    VTr1, VTr2 magnitude
    Vs+′, Vs−′, Vs+, Vs− floating supply voltages
    tHLpd high-to-low propagation delay
    tLHpd low-to-high propagation delay
    ts switching time
    tt transition time
    te establishment time
    top operating period time
    tsd shut down period time
    tlpd logic propagation delay
    tdpd driving propagation delay
    G control electrode
    S source
    AVD voltage difference area
    AH hysteresis area
    EL1, EL2 driving energy
    Φ1, Φ1′, Φ2, Φ2′ area
    RC<1 coupling ratio
    T tap
    T′ tap
    Tj junction temperature
    RC′, RC′′ RC constant

## Claims

1. A driving circuit for at least one voltage controlled power switch device comprising:
a driver signal generating circuit configured to receive a pulse width modulated signal and to generate logical pulse signals as:
a first driver signal being generated as a logical signal having a first driver signal active state only during an on-state of said pulse width modulated signal, wherein said first driver signal active state occurs at least at every rising edge of said pulse width modulated signal and wherein said first driver signal active state has a duration of at least an establishment time;
a second driver signal being generated as a logical signal having a second driver signal active state only during an off-state of said pulse width modulated signal, wherein said second driver signal active state occurs at least at every falling edge of said pulse width modulated signal and wherein said second driver signal active state has a duration of at least the establishment time, wherein a duration of the establishment time ensures a minimal driving power consumption of the driving circuit for driving the at least one voltage controlled power switch device; and

at least one trigger signal generating circuit, each trigger signal generating circuit configured to generate a trigger signal for said at least one voltage controlled power switch device, said at least one trigger signal generating circuit comprising:
a first driving transistor configured to be switched on during said first driver signal active state such that an on-voltage of the trigger signal is supplied to a control electrode of said at least one voltage controlled power switch device, and
a second driving transistor configured to be switched on during said second driver signal active state such that an off-voltage of the trigger signal is supplied to the control electrode of said at least one voltage controlled power switch device.

2. The driving circuit for at least one voltage controlled power switch device according to claim 1, comprising:
at least one energy buffer component comprising an inductivity coupled between the at least one trigger signal generating circuit and the control electrode of said at least one voltage controlled power switch device,
wherein the at least one energy buffer component is adapted to:
store signal energy of the trigger signal until a threshold voltage of said at least one voltage controlled power switch device is reached and
to release the stored signal energy to said control electrode of said at least one voltage controlled power switch device when said threshold voltage of said power switch device has been reached, and

wherein the signal energy stored in said at least one energy buffer component provides a voltage overshoot at the control electrode of said at least one voltage controlled power switch device.

3. A driving circuit for driving half bridge connected electrically controlled power switches with a near zero interlock delay time between on-states of the power switches,
wherein the driving circuit is configured to receive an input signal and to generate:
a first drive signal, wherein said first drive signal controls a first power switch and is adapted to switch between an on-voltage and an off-voltage,
a second drive signal, wherein said second drive signal controls a second power switch and is adapted to switch between said on-voltage and said off-voltage,

wherein said on-voltage is deep in an ohmic region of said first power switch and said second power switch;
wherein said off-voltage is deep in a cutoff region of said first power switch and said second power switch;
wherein transition of said first drive signal and transition of said second drive signal start simultaneously with rising and falling edges of said input signal;
wherein a signal curve of said transition of said first drive signal and a signal curve of said transition of said second drive signal have complementary slopes;
wherein the signal curve of said first drive signal is mirrored with respect to said signal curve of said second drive signal along a mirroring voltage value level at least within a transition time; and
wherein the mirroring voltage value is adjusted to be within the cutoff region of the power switches.

4. The driving circuit according to claim 3, comprising:
at least one first trigger signal generating circuit,
at least one second trigger signal generating circuit,
at least one first ramp generating circuit and
at least one second ramp generating circuit.

5. The driving circuit according to claim 4,
wherein each trigger signal generating circuit comprises a pair of triggering switches,
wherein each pair of triggering switches of the first trigger signal generating circuit is adapted to generate a first triggering signal,
wherein each pair of triggering switches of the second trigger signal generating circuit is adapted to generate a second triggering signal.

6. The driving circuit according to claim 5,
wherein as a consequence of every odd logic change of the input signal a first triggering switch of said first trigger signal generating circuit changes the value of the first triggering signal stepwise from an off-voltage state to a positive DC gate supply voltage simultaneously with a second triggering switch of the second trigger signal generating circuit changing the value of the second triggering signal stepwise from an on-voltage state to a negative DC gate supply voltage, and
wherein as a consequence of every even logic change of the input signal a first triggering switch of the second trigger signal generating circuit changes the value of the second triggering signal stepwise from an off-voltage state to a positive DC gate supply simultaneously with a second triggering switch of the first trigger signal generating circuit changing the value of the first triggering signal stepwise from an on-voltage state to the negative DC gate supply voltage.

7. The driving circuit according to claim 6,
wherein each voltage step change of the first triggering signal is shaped into the first drive signal by forming a slope via the first ramp generating circuit and wherein each voltage step change of the second triggering signal is shaped into the second drive signal by forming a slope via the second ramp generating circuit.

8. The driving circuit according to claim 7,
wherein the first ramp generating circuit comprises an energy buffer component formed by a first inductivity connected between a common point of the pair of triggering switches of the first trigger signal generating circuit and a control electrode of the first power switch and wherein the second ramp generating circuit comprises an energy buffer component formed by a second inductivity connected between a common point of the pair of the triggering switches of the second trigger signal generating circuit and a control electrode of the second power switch.

9. The driving circuit according to claim 8,
wherein the first inductivity is adapted to supply a maximum current to the control electrode of the first power switch at the time when a threshold voltage of said first power switch is reached,
wherein a first driving energy stored in the first inductivity within a switching time is high enough to form a voltage overshoot on the control electrode of the first power switch in order to accelerate the switching of the first power switch,
wherein the second inductivity is adapted to supply a maximum current to the control electrode of the second power switch at the time when the threshold voltage of the second power switch is reached and
wherein the second driving energy stored in the second inductivity within a switching time is high enough to form a voltage overshoot on the control electrode of the second power switch which accelerates the switching of the second power switch.

10. The driving circuit according to claim 9,
wherein a first damping resistor is connected in parallel to the first inductivity and is adapted to damp a signal sway of the first driving signal and
wherein a second damping resistor is connected in parallel to the second inductivity and is adapted to damp a signal sway of the second driving signal.

11. The driving circuit according to claim 4,
wherein the driving circuit comprises a gate supply switching circuit and a power supply magnetic circuit,
wherein each trigger signal generating circuit comprises a DC gate supply circuit,
wherein said DC gate supply circuit comprises at least one half wave rectifier and at least one smoothing capacitor,
wherein each DC gate supply circuit is adapted to:
generate at least one positive DC gate supply voltage and a negative DC gate supply voltage and to
supply the first triggering switch with the positive DC gate supply voltage,
supply the second triggering switch with the negative DC gate supply voltage, and to
receive energy from said power supply magnetic circuit

wherein each DC gate supply voltage can have the value 0V,
wherein said gate supply switching circuit is configured to:
receive a gate power ratio control signal,
receive a DC gate power source voltage, and to
generate a signal adapted to carry a gate supply energy;

wherein said switching circuit adjusts the voltage value of the DC gate supply voltages output by the gate supply circuit by modulating the received DC gate power source voltage with PWM information of the received gate power ratio control signal,
wherein said power supply magnetic circuit comprises at least one transformer and is adapted to provide a galvanic isolated energy transfer of said gate supply energy from the supply switching circuit to both DC gate supply circuits.

12. The driving circuit according to claim 11,
wherein the said driving circuit comprises a temperature influence adapting circuit adapted to:
receive a temperature information signal carrying out a temperature information,
generate said gate power ratio control signal,
set the value of said DC gate power source voltage via a DC/DC converter and to
set the gate power ratio control signal and the DC gate power source voltage with respect to the temperature information carried by the received temperature information signal such that the value of said DC gate supply voltages preserves a constant value of differential voltage over an entire temperature range of said temperature information signal.

13. The driving circuit according to claim 7,
wherein the mirroring between the driving signals is performed by a magnetic coupling circuit,
wherein said magnetic coupling circuit is connected between at least one first ramp generating circuit and at least one second ramp generating circuit,
wherein the magnetic coupling circuit comprises at least one transformer,
wherein said magnetic coupling circuit is adapted to perform a coupling ratio of plus one (+1) between any pair of the individual first ramp generating circuits,
wherein said magnetic coupling circuit is adapted to perform a coupling ratio of plus one (+1) between any pair of the individual second ramp generating circuits and
wherein said magnetic coupling circuit is adapted to perform a coupling ratio of minus one (−1) between said first ramp generating circuit and said second ramp generating circuit.

14. The driving circuit according to claim 13,
wherein the magnetic coupling circuit has a tap forming a coupling ratio less than one on at least one winding connected to said ramp generating circuit to form an autotransformer function,
wherein at least one of said trigger signal generating circuit comprises capping diodes connected to said tap,
wherein said magnetic coupling circuit is adapted for said coupling ratio of the magnetic coupling circuit to define a maximum voltage overshoot value on the control electrodes of the power switches and
wherein said magnetic coupling circuit and said capping diodes are adapted to return at least part of the driving energy back to the DC gate supply circuit via said capping diodes.

15. The driving circuit according to claim 3,
wherein the said driving circuit comprises a differentiating circuit and a magnetic triggering circuit having at least one transformer,
wherein the differentiating circuit is configured to receive the input signal and to generate a first and second pulsed signal,
wherein the pulsed signals both switch between an active and an inactive state and are adapted to control said pairs of said triggering switches through said magnetic triggering circuit without additional amplification of said pulsed signals,
wherein the value of the first pulsed signal is during the on state of said input signal set to an active state for an establishment time duration at least in response to each rising edge of the input signal and the value of the second pulsed signal is during the off state of the input signal set to an active state for said establishment time duration at least in response to each falling edge of said input signal,
wherein said establishment time has either a predetermined value or is defined simultaneously with respect to a power switch state information in order to ensure a minimal driving power consumption,
wherein said magnetic triggering circuit is adapted
to turn the first triggering switch of the first triggering circuit and the second triggering switch of the second triggering circuit on in response to an active state of the first pulsed signal and
to turn the second triggering switch of the first triggering circuit and the first triggering switch of the second triggering circuit on in response to an active state of the second pulsed signal and
to hold all triggering switches off if none of the pulsed signals is in an active state.

16. The driving circuit according to claim 4,
wherein said first triggering circuit comprises a bistable circuit adapted to prevent changes in the conducting state of the said power switch during the time when none of the triggering switches is switched on and
wherein said second triggering circuit comprises a bistable circuit adapted to prevent changes in the conducting state of the said power switch during the time when none of the triggering switches is switched on.

17. The driving circuit according to claim 4,
wherein at least one triggering circuit comprises a shut down circuit adapted to switch off the corresponding power switch in response to shut down information received through the signal carrying a gate supply energy.

18. The driving circuit according to claim 4,
wherein at least one of said trigger signal generating circuits comprises at least one blocking diode,
wherein said first blocking diode is connected between the DC gate supply circuit and the first triggering switch,
wherein a said second blocking diode is connected between the DC gate supply circuit and the second triggering switch.

19. The driving circuit according to claim 15,
wherein said differentiating circuit and the magnetic triggering circuit drive at least four power switches connected into a full bridge within the said driving circuit.

20. The driving circuit according to claim 3,
wherein said first power switch comprises at least one power switch element each controlled by drive signal and
wherein said second power switch comprises at least one power switch element each controlled by a drive signal.

21. A method for driving half bridge connected electrically controlled power switches in terms of having a near zero interlock delay time between on-states of the power switches comprising the steps of:
generating a first and a second triggering signal,
wherein the first triggering signal changes as a response to every odd step change of an input signal its value in a step manner from an off-voltage state to a positive DC gate supply voltage simultaneously with the second triggering signal which changes its value in a step manner from an on-voltage state to a negative DC gate supply voltage end
wherein the first triggering signal changes as a response to every even step change of an input signal its value in a step manner from an on-voltage state to a negative DC gate supply voltage simultaneously with the second triggering signal which changes its value in a step manner from an off-voltage state to a positive DC gate supply voltage,
sloping said first and said second triggering signal to a first and a second drive signal with equal slopes as a response to the rising and falling edges of the input signal,
wherein said first and second drive signal are adapted to switch a corresponding power switch among the on and off state,
wherein said first and second drive signal are mirrored with respect to one another's time curve along the time axis through a mirroring voltage value within a transition time and
wherein the said mirroring voltage value is adjusted to be low enough for said power switches to be within a cutoff region.