# Patent

## Title
Wireless communication apparatus, wireless communication method, wireless communication system, and computer program

## Abstract

There is provided a wireless communication apparatus including: a matrix estimating unit that estimates a channel matrix of N rows and M columns (N and M are natural numbers); a selecting unit that selects S rows (S is a natural number, and S<min(M, N)) from the channel matrix estimated by the matrix estimating unit and generates a sub-channel matrix of S rows and M columns; and an arithmetic unit that calculates an antenna weighting coefficient matrix based on the sub-channel matrix generated by the selecting unit. The selecting unit selects the S rows that allow the size of the sub-channel matrix to be the maximum from all matrices having N rows from which the S rows can be selected.

## Background
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to a wireless communication apparatus, a wireless communication method, a computer program, and a wireless communication system, and more particularly, to a wireless communication apparatus, a wireless communication method, a computer program, and a wireless communication system performing communication using a MIMO (multiple input multiple output) scheme.
2. Description of the Related Art
A wireless communication system has been proposed in which a transmitter and a receiver each have a plurality of antennas and space division multiplexing communication (MIMO scheme: multiple input multiple output) is performed using the plurality of antennas to increase transmission capacity.
FIG. 17 is a conceptual diagram illustrating a wireless communication system using the MIMO scheme. In a wireless communication system 1 shown in FIG. 17, a transmitter 11 includes M antennas 12a, 12b, . . . , 12m, and a receiver 21 includes N antennas 22a, 22b, . . . , 22n. 
The transmitter 11 performs space/time division multiplexing on K transmission data, distributes the data to M antennas 12a, 12b, . . . , 12m, and transmits the data through channels. Then, the receiver 21 receives the signals transmitted through the channels using N antennas 22a, 22b, 22n, and performs space/time division demultiplexing on the received signals to obtain K reception data.
Therefore, the MIMO scheme is a communication system using channel characteristics in which the transmitter 11 distributes transmission data to a plurality of antennas and transmits it through the antennas, and the receiver 21 receives signals using a plurality of antennas and processes the signals to obtain reception data.
There are various data transmission systems using the MIMO scheme. As an ideal example of the data transmission system using the MIMO scheme, an eigenmode transmission system has been known which uses the singular value decomposition (SVD) or the eigenvalue decomposition of a channel matrix.
FIG. 18 is a conceptual diagram illustrating the eigenmode transmission system. In the eigenmode transmission system using SVD, singular value decomposition is performed on a channel matrix H having channel information between transmitting and receiving antennas as elements to calculate UDVH (VH indicates a complex conjugate transposed matrix of a matrix V). Singular value decomposition is performed on the channel matrix H to calculate UDVH, the matrix V is given as a transmitter-side antenna weighting coefficient matrix, and a matrix UH (UH indicates a complex conjugate transposed matrix of a matrix U) is given as a receiver-side antenna weighting coefficient matrix. In this way, a channel can be represented by a diagonal matrix D having the square root (singular value) of an eigenvalue λi of a covariance matrix (HHH or HHH) of the channel matrix H. Therefore, it is possible to multiplex a signal and transmit the multiplexed signal, without any crosstalk.
When the number of antennas of the transmitter 11 is M, a transmission signal x′ is represented by an M×1 vector. When the number of antennas of the receiver 21 is N, a received signal y′ is represented by an N×1 vector. In addition, a channel matrix is represented as a matrix H of N rows and M columns. An element hij of the channel matrix H is a transfer function from a j-th transmitting antenna to an i-th receiving antenna (1≦i≦N and 1≦j≦M). The received signal vector y′ is obtained by adding a noise vector n to the multiplication of the channel matrix H and the transmission signal vector x′, as represented by the following Expression 1.

y′=Hx′+n  (Expression 1)

As described above, when singular value decomposition is performed on the channel matrix H, the following Expression 2 is obtained.

H=UDVH  (Expression 2)

The transmitter-side antenna weighting coefficient matrix V and the receiver-side antenna weighting coefficient matrix UH are unitary matrices that satisfy the following Expressions 3 and 4. In the following Expressions, I indicates a unitary matrix.

UHU=I  (Expression 3)

VHV=I  (Expression 4)

That is, the antenna weighting coefficient matrix UH of the receiver 21 is obtained by arranging the normalized eigenvectors of HHH, and the antenna weighting coefficient matrix V of the transmitter 11 is obtained by arranging the normalized eigenvectors of HHH. In addition, D indicates a diagonal matrix having the square root of (the singular value of H) of the eigenvalue of HHH or HHH as a diagonal component. That is, when the smaller one of the number M of transmitting antennas of the transmitter 11 and the number N of receiving antennas of the receiver 21 is referred to as L (=min(M, N)), the matrix D is a square matrix of L rows and L columns. That is, the matrix D can be represented by the following Expression 5.




  
    
      
        D
        =
        
          (
          
            
              
                
                  
                    λ
                    1
                  
                
              
              
                0
              
              
                ...
              
              
                0
              
            
            
              
                0
              
              
                
                  
                    λ
                    2
                  
                
              
              
                ...
              
              
                0
              
            
            
              
                ⋮
              
              
                ⋮
              
              
                ⋱
              
              
                ⋮
              
            
            
              
                0
              
              
                0
              
              
                ...
              
              
                
                  
                    λ
                    L
                  
                
              
            
          
          )
        
      
    
    
      
        (
        
          Expression
          ⁢
          
            
          
          ⁢
          5
        
        )
      
    
  




When the transmitter 11 multiplies data by the antenna weighting coefficient matrix V and transmits the data and the receiver 21 receives a signal and multiplies the signal by the complex conjugate transposed matrix UH, the received signal y is represented by the following Expression 6 since the matrix U of N rows and L columns and the matrix V of M rows and L columns are unitary matrices.

y=UHy′

=UH(Hx′+n)

=UH(HVx+n)

=UH(UDVH)Vx+UHn 

=(UHU)D(VHV)x+UHn 

=IDIx+UHn 

=Dx+UHn  (Expression 6)

In this case, each of the received signal y and the transmission signal x is a vector of L rows and one column. Since the matrix D is a diagonal matrix, each transmission signal transmitted from the transmitter 11 can be received by the receiver 21 without any crosstalk. Since the diagonal element of the matrix D is the square sum of the eigenvalue λi (1≦i≦L), the power of each reception signal is λi times the power of each transmission signal. In addition, for the noise component n, since the column of U is an eigenvector having a norm that is normalized to 1, UHn does not change the noise power thereof. Therefore, UHn is a vector of L rows and one column, and the received signal y and the transmission signal x have the same size.
As such, in the eigenmode transmission system using the MIMO scheme, it is possible to obtain a plurality of independent logic pulses having the same frequency at the same time without any crosstalk. That is, it is possible to wirelessly transmit a plurality of data using the same frequency at the same time, thereby improving a transmission rate.
SUMMARY OF THE INVENTION
In the MIMO transmission, an antenna weighting method in the transmitter, particularly, a weighting method for eigenmode transmission can be represented by the above-mentioned Expressions. JP-A No. 2005-160030 also discloses a weighting method using expressions.
However, in order to perform eigenmode transmission, when the number of antennas in the transmitter is M and the number of antennas in the receiver is N, computation needs to be performed on a channel matrix of N rows and M columns. In this case, in order to obtain ideal characteristics even when the number S of data streams to be transmitted is smaller than the number of antennas, computation needs to be performed on the channel matrix of N rows and M columns. In this case, the number S of data streams means the number of pulses that are actually used among the independent pulses for the eigenmode transmission, and is equal to or smaller than L.
For example, when the number of transmitting antennas is 4, the number of receiving antennas is 4, and the number of data streams is 2, it is necessary to perform computation, such as SVD, on a channel matrix H of 4 rows and 4 columns in order to perform eigenmode transmission. When singular value decomposition is performed on the channel matrix H, H=UDVH is satisfied. In this case, each of the matrices U, D, and V is a unitary matrix of 4 rows and 4 columns. In the matrix V obtained by SVD, two columns are used as an antenna weighting coefficient matrix to transmit two data streams.
In general, an excessively large amount of computation is needed to calculate the antenna weighting coefficient matrix requires, as in SVD. As the size of an input channel matrix is increased or as the rank of a matrix is increased, the amount of computation is sharply increased. When the communication system is actually used, the amount of calculation depends on the size of a circuit or a computation time. Therefore, the amount of calculation needs to be small. However, as described above, particularly, even when the number of data streams is smaller than the number of antennas, the amount of calculation depends on the number of antennas, which is not preferable in terms of the mounting of the communication system.
The present invention has been made in views of the above issues, and it is desirable to provide a wireless communication apparatus, a wireless communication method, a computer program, and a wireless communication system capable of preventing an increase in the amount of computation when the maximum value of the number of data streams to be transmitted or received is smaller than the number of antennas used to transmit data.
According to an embodiment of the present invention, there is provided a wireless communication apparatus including: a matrix estimating unit that estimates a channel matrix of N rows and M columns (N and M are natural numbers); a selecting unit that selects S rows (S is a natural number, and S<MIN(M, N)) from the channel matrix estimated by the matrix estimating unit and generates a sub-channel matrix of S rows and M columns; and an arithmetic unit that calculates an antenna weighting coefficient matrix based on the channel matrix estimated by the matrix estimating unit and the sub-channel matrix generated by the selecting unit. The arithmetic unit includes: a weighted channel matrix generating unit that multiplies the channel matrix by a temporary antenna weighting coefficient matrix calculated based on the sub-channel matrix to generate a weighted channel matrix; a reverse weighted channel matrix generating unit that transposes the channel matrix and the weighted channel matrix, and multiplies the transposed channel matrix by a reverse antenna weighting coefficient matrix calculated based on the transposed weighted channel matrix to generate a reverse weighted channel matrix; and an antenna weighting coefficient matrix calculating unit that calculates the antenna weighting coefficient matrix based on a transposed reverse weighted channel matrix.
The weighted channel matrix generating unit may repeatedly generate the weighted channel matrix using the antenna weighting coefficient matrix calculated by the antenna weighting coefficient matrix calculating unit instead of the temporary antenna weighting coefficient matrix.
The arithmetic unit may include: a first coefficient calculating unit that calculates the temporary antenna weighting coefficient matrix based on the sub-channel matrix; a first matrix multiplying unit that multiplies the channel matrix by the temporary antenna weighting coefficient matrix to generate the weighted channel matrix; a first transposition unit that transposes the channel matrix and the weighted channel matrix; a second coefficient calculating unit that calculates the reverse antenna weighting coefficient matrix based on the transposed weighted channel matrix; a second matrix multiplying unit that multiplies the transposed channel matrix by the reverse antenna weighting coefficient matrix to generate the reverse weighted channel matrix; a second transposition unit that transposes the reverse weighted channel matrix; and a third coefficient calculating unit that calculates the antenna weighting coefficient matrix based on the transposed reverse weighted channel matrix.
The first coefficient calculating unit may calculate the temporary antenna weighting coefficient matrix based on the sub-channel matrix and the channel matrix. The second coefficient calculating unit may calculate the reverse antenna weighting coefficient matrix based on the transposed weighted channel matrix and the transposed channel matrix. The third coefficient calculating unit may calculate the antenna weighting coefficient matrix based on the transposed reverse weighted channel matrix and the channel matrix.
At least one of the first to third coefficient calculating units may include: a fourth coefficient calculating unit that calculates the temporary antenna weighting coefficient matrix based on the sub-channel matrix; a third matrix multiplying unit that multiplies the channel matrix by the temporary antenna weighting coefficient matrix to generate the weighted channel matrix; a singular value decomposing unit that performs singular value decomposition on the weighted channel matrix; and a fourth matrix multiplying unit that multiplies a matrix obtained by the singular value decomposition by the temporary antenna weighting coefficient matrix to generate the antenna weighting coefficient matrix.
The selecting unit may select the S rows that allow the sum of the squares of elements of a matrix to be the maximum. The selecting unit may select the S rows that allow an eigenvalue of a covariance matrix of a matrix or the maximum value of a singular value of the matrix to be the maximum. The selecting unit may select the S rows that allow an eigenvalue of a covariance matrix of a matrix or the minimum value of a singular value of the matrix to be the maximum. The selecting unit may select the S rows that allow a determinant of a covariance matrix of a matrix to be the maximum. The selecting unit may select the S rows that allow a value obtained by dividing a determinant of a covariance matrix of a matrix by the sum of the squares of elements of the matrix to be the maximum.
According to another embodiment of the present invention, there is provided a wireless communication method including the steps of: estimating a channel matrix of N rows and M columns (N and M are natural numbers); selecting S rows (S is a natural number, and S<MIN(M, N)) from the estimated channel matrix and generating a sub-channel matrix of S rows and M columns; and calculating an antenna weighting coefficient matrix based on the estimated channel matrix and the generated sub-channel matrix. The calculating step includes the sub-steps of: multiplying the channel matrix by a temporary antenna weighting coefficient matrix calculated based on the sub-channel matrix to generate a weighted channel matrix; transposing the channel matrix and the weighted channel matrix, and multiplying the transposed channel matrix by a reverse antenna weighting coefficient matrix calculated based on the transposed weighted channel matrix to generate a reverse weighted channel matrix; and calculating the antenna weighting coefficient matrix based on a transposed reverse weighted channel matrix.
According to another embodiment of the present invention, there is provided a computer program for allowing a computer to execute the steps of: estimating a channel matrix of N rows and M columns (N and M are natural numbers); selecting S rows (S is a natural number, and S<MIN(M, N)) from the estimated channel matrix and generating a sub-channel matrix of S rows and M columns; and calculating an antenna weighting coefficient matrix based on the estimated channel matrix and the generated sub-channel matrix. The calculating-step includes the sub-steps of: multiplying the channel matrix by a temporary antenna weighting coefficient matrix calculated based on the sub-channel matrix to generate a weighted channel matrix; transposing the channel matrix and the weighted channel matrix, and multiplying the transposed channel matrix by a reverse antenna weighting coefficient matrix calculated based on the transposed weighted channel matrix to generate a reverse weighted channel matrix; and calculating the antenna weighting coefficient matrix based on a transposed reverse weighted channel matrix.
According to another embodiment of the present invention, there is provided a wireless communication system including: a first wireless communication apparatus; and a second wireless communication apparatus. The first wireless communication apparatus includes a transmitting unit that transmits reference signals to the second wireless communication apparatus through a plurality of antennas. The second wireless communication apparatus includes: a matrix estimating unit that estimates a channel matrix of N rows and M columns (N and M are natural numbers) based on the reference signals received through a plurality of antennas; a selecting unit that selects S rows (S is a natural number, and S<MIN(M, N)) from the channel matrix estimated by the matrix estimating unit and generates a sub-channel matrix of S rows and M columns; and an arithmetic unit that calculates an antenna weighting coefficient matrix based on the channel matrix estimated by the matrix estimating unit and the sub-channel matrix generated by the selecting unit. The arithmetic unit includes: a weighted channel matrix generating unit that multiplies the channel matrix by a temporary antenna weighting coefficient matrix calculated based on the sub-channel matrix to generate a weighted channel matrix; a reverse weighted channel matrix generating unit that transposes the channel matrix and the weighted channel matrix, and multiplies the transposed channel matrix by a reverse antenna weighting coefficient matrix calculated based on the transposed weighted channel matrix to generate a reverse weighted channel matrix; and an antenna weighting coefficient matrix calculating unit that calculates the antenna weighting coefficient matrix based on a transposed reverse weighted channel matrix.
As described above, according to the above-mentioned embodiments of the present invention, it is possible to provide a wireless communication apparatus, a wireless communication method, a computer program, and a wireless communication system capable of preventing an increase in the amount of computation when the maximum value of the number of data streams to be transmitted or received is smaller than the number of antennas used to transmit data.



BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a diagram illustrating a wireless communication system 10 according to a first embodiment of the present invention;
FIG. 2 is a diagram illustrating the structure of a weighting coefficient matrix calculating circuit 104 according to the first embodiment of the present invention;
FIG. 3 is a diagram illustrating the structure of an arithmetic circuit 114 according to the first embodiment of the present invention;
FIG. 4 is a flowchart illustrating a method of calculating an antenna weighting coefficient matrix W according to the first embodiment of the present invention;
FIG. 5A is a line graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the first embodiment of the present invention;
FIG. 5B is a line graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the first embodiment of the present invention;
FIG. 5C is a line graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the first embodiment of the present invention;
FIG. 5D is a line graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the first embodiment of the present invention;
FIG. 5E is a line graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the first embodiment of the present invention;
FIG. 5F is a line graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the first embodiment of the present invention;
FIG. 6A is a bar graph illustrating the convergence of communication capacity;
FIG. 6B is a bar graph illustrating the convergence of communication capacity;
FIG. 6C is a bar graph illustrating the convergence of communication capacity;
FIG. 6D is a bar graph illustrating the convergence of communication capacity;
FIG. 6E is a bar graph illustrating the convergence of communication capacity;
FIG. 6F is a bar graph illustrating the convergence of communication capacity;
FIG. 7 is a diagram illustrating the structure of an arithmetic circuit 214 according to a second embodiment of the present invention;
FIG. 8 is a flowchart illustrating a method of calculating an antenna weighting coefficient matrix W according to the second embodiment of the present invention;
FIG. 9 is a bar graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the second embodiment of the present invention;
FIG. 10A is a bar graph illustrating the ratio of communication capacities;
FIG. 10B is a bar graph illustrating the ratio of communication capacities;
FIG. 10C is a bar graph illustrating the ratio of communication capacities;
FIG. 10D is a bar graph illustrating the ratio of communication capacities;
FIG. 10E is a bar graph illustrating the ratio of communication capacities;
FIG. 10F is a bar graph illustrating the ratio of communication capacities;
FIG. 11 is a diagram illustrating the structure of an arithmetic circuit 414 according to a third embodiment of the present invention;
FIG. 12 is a diagram illustrating the structure of a first coefficient calculating circuit 422a according to the third embodiment of the present invention;
FIG. 13 is a flowchart illustrating a method of calculating an antenna weighting coefficient matrix W according to the third embodiment of the present invention;
FIG. 14A is a line graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the third embodiment of the present invention;
FIG. 14B is a line graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the third embodiment of the present invention;
FIG. 14C is a line graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the third embodiment of the present invention;
FIG. 14D is a line graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the third embodiment of the present invention;
FIG. 14E is a line graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the third embodiment of the present invention;
FIG. 14F is a line graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the third embodiment of the present invention;
FIG. 15 is a diagram illustrating the structure of an arithmetic circuit 414′ according to a modification of the third embodiment of the present invention;
FIG. 16A is a bar graph illustrating the convergence of communication capacity;
FIG. 16B is a bar graph illustrating the convergence of communication capacity;
FIG. 16C is a bar graph illustrating the convergence of communication capacity;
FIG. 16D is a bar graph illustrating the convergence of communication capacity;
FIG. 16E is a bar graph illustrating the convergence of communication capacity;
FIG. 16F is a bar graph illustrating the convergence of communication capacity;
FIG. 17 is a conceptual diagram illustrating a wireless communication system using a MIMO scheme; and
FIG. 18 is a conceptual diagram illustrating eigenmode transmission.



DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
Hereinafter, preferred embodiments of the present invention will be described in detail with reference to the appended drawings. Note that, in this specification and the appended drawings, structural elements that have substantially the same function and structure are denoted with the same reference numerals, and repeated explanation of these structural elements is omitted.
First Embodiment
First, a wireless communication apparatus and a wireless communication system using the wireless communication apparatus according to a first embodiment of the present invention will be described.
FIG. 1 is a diagram illustrating a wireless communication system 10 according to the first embodiment of the present invention. Hereinafter, the wireless communication system 10 according to the first embodiment of the present invention will be described with reference to FIG. 1.
As shown in FIG. 1, the wireless communication system 10 according to the first embodiment of the present invention includes a transmitter 100 and a receiver 200.
Each of the transmitter 100 and the receiver 200 is an example of the wireless communication apparatus according to the present invention. The transmitter 100 transmits data to the receiver 200 using a plurality of antennas by a MIMO scheme, and the receiver 200 receives data transmitted from the transmitter 100 using a plurality of antennas by the MIMO scheme.
A data link from the transmitter 100 to the receiver 200 is defined as a forward link, and a link from the receiver 200 to the transmitter 100 is defined as a backward link. A channel matrix H of the forward link is a matrix of N rows and M columns, and a channel matrix HB of the backward link is a matrix of M rows and N columns. In addition, ideally, the channel matrix satisfies the following relationship represented by Expression 7.

HB=HT  (Expression 7)

The transmitter 100 includes M antennas 101a, 101b, . . . , 101m, a channel estimating circuit 102, a weighting coefficient matrix calculating circuit 104, and a transmitting circuit 106. The transmitting circuit 106 includes a weighting circuit 110.
The channel estimating circuit 102 estimates a channel matrix HB from the receiver 200 to the transmitter 100 using a known pattern (for example, a reference signal) transmitted from the receiver 200. A known method can be used to estimate the channel matrix, and thus a detailed description thereof will be omitted. Ideally, HB=HT is satisfied by Expression 7. Therefore, it is possible to estimate a channel matrix H using the channel estimating circuit 102. The channel matrix H estimated by the channel estimating circuit 102 is input to the weighting coefficient matrix calculating circuit 104.
The weighting coefficient matrix calculating circuit 104 uses the channel matrix H transmitted from the channel estimating circuit 102 to calculate an antenna weighting coefficient matrix W for a transmission beam-forming. The antenna weighting coefficient matrix W generated by the weighting coefficient matrix calculating circuit 104 is input to the weighting circuit 110.
The transmitting circuit 106 receives a transmission data stream transmitted from the transmitter 100 to the receiver 200, and generates transmission signals to be transmitted from the antennas 101a, 101b, . . . , 101m. The weighting circuit 110 multiplies a transmission data stream x transmitted from the transmitter 100 to the receiver 200 by the antenna weighting coefficient matrix W generated by the weighting coefficient matrix calculating circuit 104 to generate a transmission signal x′. The transmission signal x′ generated by the weighting circuit 110 is radiated from each of the antennas, thereby performing transmission beam-forming.
The channel estimating circuit 102 may be provided in the receiver 200. When the channel estimating circuit 102 is provided in the receiver 200, it receives a known pattern transmitted from the transmitter 100, and estimates the channel matrix H based on the received pattern. The receiver 200 notifies the channel matrix H to the transmitter 100 using any method (for example, a method of transmitting data related to the channel matrix H from the receiver 200 to the transmitter 100).
The weighting coefficient matrix calculating circuit 104 may also be provided in the receiver 200. When the weighting coefficient matrix calculating circuit 104 is provided in the receiver 200, it calculates the antenna weighting coefficient matrix W based on the channel matrix H. Then, the receiver 200 notifies the antenna weighting coefficient matrix W to the transmitter 100 using any method (for example, a method of transmitting data related to the antenna weighting coefficient matrix W from the receiver 200 to the transmitter 100).
The transmitting circuit 202 generates transmission signals to be transmitted from the receiver 200 to the transmitter through the antennas 201a, 201b, . . . , 201n. The transmitting circuit 202 transmits a known pattern (for example, a reference signal) used for the channel estimating circuit 102 to estimate the channel matrix H to the transmitter 100 through the antennas 201a, 201b, . . . , 201n. 
The wireless communication system 10 according to the first embodiment of the present invention has been described above with reference to FIG. 1. Next, the structure of the weighting coefficient matrix calculating circuit 104 according to the first embodiment of the present invention will be described.
FIG. 2 is a diagram illustrating the structure of the weighting coefficient matrix calculating circuit 104 according to the first embodiment of the present invention. Hereinafter, the structure of the weighting coefficient matrix calculating circuit according to the first embodiment of the present invention will be described with reference to FIG. 2.
As shown in FIG. 2, the weighting coefficient matrix calculating circuit 104 according to the first embodiment of the present invention includes a selecting circuit 112 and an arithmetic circuit 114.
The selecting circuit 112 appropriately selects S rows from an input channel matrix H of N rows and M columns, and outputs a sub-channel matrix HS. Here, S indicates the maximum value of the number of data streams, and S, N, and M satisfy S<min(M, N). The selecting circuit 112 outputs the sub-channel matrix HS of S rows and M columns. The sub-channel matrix HS output from the selecting circuit 112 is transmitted to the arithmetic circuit 114.
The arithmetic circuit 114 calculates the antenna weighting coefficient matrix W for a transmission beam-forming based on the channel matrix H estimated by the channel estimating circuit 102 and the sub-channel matrix HS transmitted from the selecting circuit 112 and outputs the calculated antenna weighting coefficient matrix W. The antenna weighting coefficient matrix W output from the arithmetic circuit 114 is transmitted to the weighting circuit 110. A method of calculating the antenna weighting coefficient matrix W based on the sub-channel matrix HS in the arithmetic circuit 114 will be described in detail below.
The structure of the weighting coefficient matrix calculating circuit 104 according to the first embodiment of the present invention has been described above with reference to FIG. 2. Next, the structure of the arithmetic circuit 114 according to the first embodiment of the present invention will be described.
FIG. 3 is a diagram illustrating the structure of the arithmetic circuit 114 according to the first embodiment of the present invention. Hereinafter, the structure of the arithmetic circuit 114 according to the first embodiment of the present invention will be described with reference to FIG. 3.
As shown in FIG. 3, the arithmetic circuit 114 according to the first embodiment of the present invention includes coefficient calculating circuits 322a, 322b, and 322c, matrix multiplying circuits 324a and 324b, and transposition circuits 326a, 326b, and 326c. 
The coefficient calculating circuit 322a calculates a temporary antenna weighting coefficient matrix WT based on the sub-channel matrix HS. For example, the coefficient calculating circuit 322a can use singular value decomposition to calculate the temporary antenna weighting coefficient matrix WT. When singular value decomposition is performed on the sub-channel matrix HS, the above-mentioned Expression 8 is obtained.

HS=UDVH  (Expression 8)

When the temporary antenna weighting coefficient matrix WT is calculated by singular value decomposition, the coefficient calculating circuit 322a uses the matrix V (right singular matrix) of the Expression 8 as the temporary antenna weighting coefficient matrix WT. The coefficient calculating circuit 322a outputs the calculated temporary antenna weighting coefficient matrix WT to the matrix multiplying circuit 324a. 
The matrix multiplying circuit 324a is for multiplying matrices. The matrix multiplying circuit 324a according to this embodiment multiplies the channel matrix H estimated by the channel estimating circuit 102 by the temporary antenna weighting coefficient matrix WT calculated by the coefficient calculating circuit 322a to generate a weighted channel matrix HW(HW=HWT). In this case, HW indicates a matrix of M rows and S columns. The weighted channel matrix HW generated by the matrix multiplying circuit 324a is output to the transposition circuit 326a. 
The transposition circuit 326a is for transposing a matrix. The transposition circuit 326a according to this embodiment receives the weighted channel matrix HW generated by the matrix multiplying circuit 324a, transposes the weighted channel matrix HW, and outputs a transposed weighted channel matrix HWT. The transposed weighted channel matrix HWT is input to the coefficient calculating circuit 322b. 
The coefficient calculating circuit 322b calculates a reverse antenna weighting coefficient matrix WR based on the transposed weighted channel matrix HWT output from the transposition circuit 326a. For example, the coefficient calculating circuit 322b can use singular value decomposition to calculate the reverse antenna weighting coefficient matrix WR. When singular value decomposition is performed on the transposed weighted channel matrix HWT, the following Expression 9 is obtained.

HWT=UDVH  (Expression 9)

When the reverse antenna weighting coefficient matrix WR is calculated by singular value decomposition, the coefficient calculating circuit 322b uses the matrix (right singular matrix) V of Expression 13 as the reverse antenna weighting coefficient matrix WR. The coefficient calculating circuit 322b outputs the calculated reverse antenna weighting coefficient matrix WR to the matrix multiplying circuit 324b. 
The transposition circuit 326b is for transposing a matrix. The transposition circuit 326b according to this embodiment receives the channel matrix H estimated by the channel estimating circuit 102, transposes the channel matrix H, and outputs a transposed channel matrix HT. The transposed channel matrix HT is input to the matrix multiplying circuit 324b. 
The matrix multiplying circuit 324b is for multiplying matrices. The matrix multiplying circuit 324b according to this embodiment multiplies the reverse antenna weighting coefficient matrix WR output from the coefficient calculating circuit 322b by the transposed channel matrix HT output from the transposition circuit 326b to generate a reverse weighted channel matrix HWR (HWR=HTWR). In the case, the reverse weighted channel matrix HWR includes N rows and S columns. The reverse weighted channel matrix HWR generated by the matrix multiplying circuit 324b is input to the transposition circuit 326c. 
The transposition circuit 326c is for transposing a matrix. The transposition circuit 326c according to this embodiment transposes the reverse weighted channel matrix HWR generated by the matrix multiplying circuit 324b to generate a transposed reverse weighted channel matrix HWRT. The transposed reverse weighted channel matrix HWRT is input to the coefficient calculating circuit 322c. 
The coefficient calculating circuit 322c receives the transposed reverse weighted channel matrix HWRT output from the transposition circuit 326c, and generates the antenna weighting coefficient matrix W based on the transposed reverse weighted channel matrix HWRT. For example, the coefficient calculating circuit 322c can use singular value decomposition to calculate the antenna weighting coefficient matrix W. When singular value decomposition is performed on the transposed reverse weighted channel matrix HWRT, the following Expression 14 is obtained.

HWRT=UDVH  (Expression 10)

When the antenna weighting coefficient matrix W is calculated by singular value decomposition, the coefficient calculating circuit 322c uses the matrix (right singular matrix) V of Expression 10 as the antenna weighting coefficient matrix W. The coefficient calculating circuit 322c outputs the calculated antenna weighting coefficient matrix W to the weighting circuit 110.
The antenna weighting coefficient matrix W output from the coefficient calculating circuit 322c may be input to the matrix multiplying circuit 324a again. It is possible to improve communication characteristics by repeatedly performing a computing process using the antenna weighting coefficient matrix W output from the coefficient calculating circuit 322c. 
The structure of the arithmetic circuit 114 according to the first embodiment of the present invention has been described above with reference to FIG. 3. Next, a method of calculating the antenna weighting coefficient matrix W according to the first embodiment of the present invention will be described.
FIG. 4 is a flowchart illustrating the method of calculating the antenna weighting coefficient matrix W according to the first embodiment of the present invention. Hereinafter, the method of calculating the antenna weighting coefficient matrix W according to the first embodiment of the present invention will be described with reference to FIG. 4.
When the channel matrix H of N rows and M columns is input to the weighting coefficient matrix calculating circuit 104, the selecting circuit 112 appropriately selects S rows from the channel matrix H and generates the sub-channel matrix HS (Step S302). When the selecting circuit 112 generates the sub-channel matrix HS, the coefficient calculating circuit 322a calculates the temporary antenna weighting coefficient matrix WT based on the generated sub-channel matrix HS (Step S304). As described above, for example, the coefficient calculating circuit 322a can use singular value decomposition indicated in Expression 8 to calculate the temporary antenna weighting coefficient matrix WT.
When the temporary antenna weighting coefficient matrix WT is completely generated in Step S304, the matrix multiplying circuit 324a multiplies the channel matrix H by the temporary antenna weighting coefficient matrix WT to generate the weighted channel matrix HW (Step S306).
When the weighted channel matrix HW is generated in Step S306, the transposition circuit 326a transposes the weighted channel matrix HW, and the coefficient calculating circuit 322b performs singular value decomposition on the transposed weighted channel matrix HWT to generate the reverse antenna weighting coefficient matrix WR (Step S308).
When the reverse antenna weighting coefficient matrix WR is generated in Step S308, the matrix multiplying circuit 324b multiplies the transposed channel matrix HT transposed from the channel matrix H by the transposition circuit 326b by the reverse antenna weighting coefficient matrix WR. The reverse weighted channel matrix HWR is obtained by the multiplication of the transposed channel matrix HT and the reverse antenna weighting coefficient matrix WR (Step S310).
When the reverse weighted channel matrix HWR is obtained in Step S310, the transposition circuit 326c transposes the reverse weighted channel matrix HWR to obtain the transposed reverse weighted channel matrix HWRT. Then, the coefficient calculating circuit 322c performs singular value decomposition on the transposed reverse weighted channel matrix HWRT to calculate the antenna weighting coefficient matrix W (Step 312).
In this way, finally, the matrix is transposed to obtain the antenna weighting coefficient matrix W. Therefore, it is possible to obtain the same effects as those when the arithmetic circuit 114 performs a two-way beam-forming process.
From the relationship of S<min(M, N), since the rank of the matrix subjected to singular value decomposition is less than that in eigenmode transmission, the amount of calculation is reduced by a value corresponding to the reduction in the rank of the matrix. In addition, the amount of calculation is reduced by a value corresponding to a reduction in the number of elements of the matrix generated by the singular value decomposition. In addition, in order to improve communication characteristics, Step S306 to Step S312 may be repeatedly performed using the antenna weighting coefficient matrix W output from the coefficient calculating circuit 322c. As described above, the total number of processes is increased, but the amount of calculation of the complicated singular value decomposition process is reduced. Therefore, it is possible to prevent an increase in the total amount of computation.
The coefficient calculating circuits 322a, 322b, and 322c may be integrated with each other, the matrix multiplying circuits 324a, 324b, and 324c may be integrated with each other, and the transposition circuits 326a, 326b, and 326c may be integrated with each other according to conditions.
FIGS. 5A to 5F are line graphs illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the first embodiment of the present invention. FIGS. 5A to 5F plot the maximum value and the minimum value of the ratios of communication capacity C1 when S data streams are transmitted by the antenna weighting coefficient matrix W to communication capacity C0 when S data streams are transmitted by eigenmode transmission (C1/C0) for 10000 samples. In the line graphs, the horizontal axis indicates the number of times Steps S306 to S312 are repeated. In addition, the number of repetitions is zero when the temporary antenna weighting matrix WT calculated by the coefficient calculating circuit 322a is used instead of the antenna weighting coefficient matrix W.
The communication capacity C is calculated by Expression 11 given below.

C=log2(det(I+HWHHW·P/(Mσ2)))  (Expression 11)

P indicates transmission power, and is 1 in Expression 11. σ2 indicates noise power and is 0.1 (SNR=10 dB) in Expression 11. In addition, M indicates the number of transmitting antennas, and I indicates a unitary matrix of S rows and S columns. HW indicates a weighted channel matrix, and is represented by the following. Expression 12. In addition, it is assumed that each element of the channel matrix H has a complex normal distribution with a mean of 0 and a variance of 1.

HW=HW  (Expression 12)

In FIGS. 5A to 5F, when the number of transmitting antennas is M, the number of receiving antennas is N, and the number of rows selected from the channel matrix is S, the ratio of communication capacities is represented by six combinations of M, N, and S.
FIG. 5A is a line graph illustrating characteristics when (M×N×S) is (3×3×2), FIG. 5B is a line graph illustrating characteristics when (M×N×S) is (4×4×2), and FIG. 5C is a line graph illustrating characteristics when (M×N×S) is (4×4×3). In addition, FIG. 5D is a line graph illustrating characteristics when (M×N×S) is (5×5×2), FIG. 5E is a line graph illustrating characteristics when (M×N×S) is (5×5×3), and FIG. 5F is a line graph illustrating characteristics when (M×N×S) is (5×5×4).
As can be seen from six line graphs shown in FIGS. 5A to 5F, the ratio of the communication capacities converges on 1. That is, the communication capacity when the antenna weighting coefficient matrix W is used converges on the communication capacity during the eigenmode transmission.
Then, in order to improve the communication capacity, the selecting circuit 112 uses the following patterns. In the channel matrix H of N rows and M columns, the number of matrices capable of selecting S rows from N rows is NCS. A set of NCS matrices is referred to as G(H).
When one matrix H0 (H0 is a matrix of S rows and M columns) is selected from G(H), for example, a large matrix may be selected, and a matrix that is easy to calculate its inverse matrix (a matrix that is ease to calculate its inverse matrix is defined as a matrix having high reversibility) may be selected. In this embodiment, when one matrix H0 is selected from G(H), the following five patterns are used. However, the present invention is not limited to the example in which S rows are selected from N rows.
(Pattern 1)
A matrix whose covariance matrix has the largest minimum value of its eigenvalue (or its singular value) is selected from G(H). That is, when H0εG(H) and the minimum value of the eigenvalue of H0HH0 or H0H0H (or the singular value of H0) is λmin, a matrix H0 having the largest value of λmin is selected, and is output as the sub-channel matrix HS.
(Pattern 2)
A matrix whose covariance matrix has the largest determinant is selected from G(H). That is, a matrix H0 that satisfies H0εG(H) and has the maximum value of det(H0HH0) is selected and is output as the sub-channel matrix HS. H0H indicates a complex conjugate transposed matrix of the matrix H0.
(Pattern 3)
A matrix whose covariance matrix has the largest maximum value of its eigenvalue (or its singular value) is selected from G(H). That is, when H0εG(H) and the maximum value of the eigenvalue of H0HH0 or H0H0H (or the singular value of H0) is λmax, a matrix H0 having the maximum value of λmax is selected, and is output as the sub-channel matrix HS.
(Pattern 4)
A matrix in which the sum of the squares of its elements is the maximum is selected from G(H). That is, a matrix H0 that satisfies H0εG(H) and has the maximum value of trace(H0HH0) is selected and is output as the sub-channel matrix HS.
(Pattern 5)
A matrix that has the maximum value when the determinant of its covariance matrix is divided by the sum of the squares of its elements is selected from G(H). That is, a matrix H0 that satisfies H0εG(H) and has the maximum value of det(H0HH0)/trace(H0HH0) is selected and is output as the sub-channel matrix HS. In the pattern 5, a matrix H0 having the maximum value of det(H0HH0)α/trace(H0HH0)β (α and β are arbitrary values) may be selected.
A pattern that selects a large matrix corresponds to the patterns 3 and 4, and a pattern that selects a matrix having the highest reversibility corresponds to the patterns 1, 2, and 5.
FIGS. 6A to 6F are line graphs illustrating the convergence of communication capacity when the antenna weighting coefficient matrix W is calculated using the sub-channel matrix HS selected by the patterns 1 to 5. FIGS. 6A to 6F show the antenna weighting coefficient matrix W calculated using the sub-channel matrix HS that is selected by the patterns 1 to 5 when (M×N×S) is (3×3×2).
FIG. 6A is the same as FIG. 5A, FIG. 6B is a line graph illustrating characteristics when the sub-channel matrix HS selected by the pattern 1 is used, and FIG. 6C is a line graph illustrating characteristics when the sub-channel matrix HS selected by the pattern 2 is used. FIG. 6D is a line graph illustrating characteristics when the sub-channel matrix HS selected by the pattern 3 is used, FIG. 6E is a line graph illustrating characteristics when the sub-channel matrix HS selected by the pattern 4 is used, and FIG. 6F is a line graph illustrating characteristics when the sub-channel matrix HS selected by the pattern 5 is used.
As can be seen from the graphs shown in FIGS. 6A to 6F, in all the patterns 1 to 5, convergence characteristics are improved, as compared to when S rows are appropriately selected from N rows of the channel matrix H of N rows and M columns to generate the sub-channel matrix HS. The improvement in the convergence characteristics makes it possible to shorten a calculation time.
In this embodiment, (M×N×S) is (3×3×2), but other combinations of M, N, and S may be used. In this case, the convergence characteristics are also improved, as compared to when S rows are appropriately selected from N rows of the channel matrix H of N rows and M columns to generate the sub-channel matrix HS.
The method of calculating the antenna weighting coefficient matrix W according to the first embodiment of the present invention has been described above with reference to FIG. 4. As described above, according to the first embodiment of the present invention, when the maximum value of the number of data streams to be subjected to a transmission beam-forming process is smaller than the number of antennas, the antenna weighting coefficient matrix W is calculated based on the channel matrix H and the sub-channel matrix HS obtained by extracting the number of rows corresponding to the maximum value of the number of data streams from the channel matrix H.
As such, since the antenna weighting coefficient matrix W is calculated based on the channel matrix H and the sub-channel matrix HS, the rank of the matrix used for calculation is reduced, and it is possible to reduce the amount of calculation of SVD, which is a representative example of the matrix computation, while maintaining characteristics at a certain level. In addition, it is possible to prevent deterioration of characteristics and reduce the overall size of a weighting coefficient matrix calculating circuit. When this wireless communication system is actually mounted to a communication apparatus, the amount of computation depends on the size of a circuit and a computation time. Therefore, it is possible to reduce the size of a circuit and shorten the computation time.
When the sub-channel matrix HS is generated, it is possible to improve characteristics while minimizing an increase in the amount of computation by giving row selection conditions.
Second Embodiment
In the first embodiment of the present invention, after the temporary antenna weighting coefficient matrix WT is calculated from the sub-channel matrix HS, the reverse antenna weighting coefficient matrix WR is obtained by the transposition of the matrix, and the antenna weighting coefficient matrix W is obtained by the transposition of the matrix. In addition, it is possible to improve the convergence characteristics of communication capacity by changing the structures of the coefficient calculating circuits 322a, 322b, and 322c shown in FIG. 3. In a second embodiment of the present invention, the antenna weighting coefficient matrix W is calculated using the channel matrix H and the sub-channel matrix HS, in order to achieve the improvement of the convergence characteristics, which will be described below.
FIG. 7 is a diagram illustrating the structure of the arithmetic circuit 214 according to the second embodiment of the present invention. Hereinafter, the structure of the arithmetic circuit 214 according to the second embodiment of the present invention will be described with reference to FIG. 7.
The arithmetic circuit 214 shown in FIG. 7 replaces the arithmetic circuit 114 according to the first embodiment of the present invention shown in FIG. 2. That is, the arithmetic circuit 214 receives the sub-channel matrix HS selected by the selecting circuit 112 and the channel matrix H estimated by the channel estimating circuit 102, and outputs the antenna weighting coefficient matrix W for a transmission beam-forming. As shown in FIG. 7, the arithmetic circuit 214 according to the second embodiment of the present invention includes a coefficient calculating circuit 222, matrix multiplying circuits 224a and 224b, and an SVD circuit 226.
The coefficient calculating circuit 222 calculates a temporary antenna weighting coefficient matrix WT based on the sub-channel matrix HS. For example, singular value decomposition may be used as a method of calculating the temporary antenna weighting coefficient matrix WT in the coefficient calculating circuit 222. When singular value decomposition is performed on the sub-channel matrix HS, the following Expression 13 is obtained.

HS=UDVH  (Expression 13)

When the singular value decomposition is performed to calculate the temporary antenna weighting coefficient matrix WT, the coefficient calculating circuit 222 uses the matrix (right singular matrix) V of Expression 13 as the temporary antenna weighting coefficient matrix WT. The coefficient calculating circuit 222 outputs the calculated temporary antenna weighting coefficient matrix WT to the matrix multiplying circuits 224a and 224b. 
The matrix multiplying circuit 224a is for multiplying matrices. The matrix multiplying circuit 224a according to this embodiment multiplies the channel matrix H estimated by the channel estimating circuit 102 by the temporary antenna weighting coefficient matrix WT calculated by the coefficient calculating circuit 222 to generate a weighted channel matrix HW (HW=HT). The matrix HW includes M rows and S columns. The weighted channel matrix HW generated by the matrix multiplying circuit 224a is output to the SVD circuit 226.
The SVD circuit 226 is for performing singular value decomposition. In this embodiment, the SVD circuit 226 receives the weighted channel matrix HW generated by the matrix multiplying circuit 224a and performs the singular value decomposition on the weighted channel matrix HW. When the singular value decomposition is performed on the weighted channel matrix HW, the following Expression 14 is obtained.

HW=UDVH  (Expression 14)

The matrix (right singular matrix) V obtained by the Expression 14 is output to the matrix multiplying circuit 224b. 
The matrix multiplying circuit 224b is for multiplying matrices. The matrix multiplying circuit 224b according to this embodiment multiplies the temporary antenna weighting coefficient matrix WT calculated by the coefficient calculating circuit 222 by the matrix V calculated by the SVD circuit 226 to generate the antenna weighting coefficient matrix W (W=WTV).
The structure of the arithmetic circuit 214 according to the second embodiment of the present invention has been described with reference to FIG. 7. Next, a method of calculating the antenna weighting coefficient matrix W according to the second embodiment of the present invention will be described.
FIG. 8 is a flowchart illustrating the method of calculating the antenna weighting coefficient matrix W according to the second embodiment of the present invention. Hereinafter, the method of calculating the antenna weighting coefficient matrix W according to the second embodiment of the present invention will be described with reference to FIG. 8.
When a channel matrix H of N rows and M columns is input to the weighting coefficient matrix calculating circuit 104, the selecting circuit 112 appropriately selects S rows from the channel matrix H and generates a sub-channel matrix Hs (Step S202). When the selecting circuit 112 generates the sub-channel matrix HS, the coefficient calculating circuit 222 calculates the temporary antenna weighting coefficient matrix WT based on the generated sub-channel matrix HS (Step S204). As described above, for example, the coefficient calculating circuit 222 can use singular value decomposition as indicated in Expression 13 to calculate the temporary antenna weighting coefficient matrix WT.
When the temporary antenna weighting coefficient matrix WT is completely generated in Step S204, the matrix multiplying circuit 224a multiplies the channel matrix H by the temporary antenna weighting coefficient matrix WT to generate the weighted channel matrix HW (Step S206). As such, the weighted channel matrix HW is generated, and an antenna weighting coefficient matrix most suitable for the channel matrix H is calculated again.
When the weighted channel matrix HW is generated in Step S206, the SVD circuit 226 performs singular value decomposition on the weighted channel matrix HW to obtain the matrix V (Step S208).
When the singular value decomposition is performed on the weighted channel matrix HW in Step S208, the matrix multiplying circuit 224b multiplies the temporary antenna weighting coefficient matrix WT generated in Step S204 by the matrix V obtained in Step S208 to calculate the antenna weighting coefficient matrix W (Step S210). The rows that are not used in the calculation of the temporary antenna weighting coefficient matrix WT (that is, the rows of the channel matrix H that are not included in the sub-channel matrix HS) are used to calculate the antenna weighting coefficient matrix W. Therefore, it is possible to improve characteristics.
From the relationship of S<min(M, N), since the rank of the matrix subjected to singular value decomposition is less than that in eigenmode transmission, the amount of calculation is reduced by a value corresponding to the reduction in the rank of the matrix. In addition, the amount of calculation is reduced by a value corresponding to a reduction in the number of elements of the matrix generated by the singular value decomposition. As described above, the total number of processes is increased, but the amount of calculation of the complicated singular value decomposition process is reduced. Therefore, it is possible to prevent an increase in the total amount of computation.
In addition, the coefficient calculating circuit 222 and the SVD circuit 226 may be integrated with each other according to conditions.
FIG. 9 is a bar graph illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the second embodiment of the present invention. FIG. 9 plots the average value of the ratios of communication capacity C1 when S data streams are transmitted by using the antenna weighting coefficient matrix W to communication capacity C0 when S data streams are transmitted by eigenmode transmission (C1/C0) for 10000 samples. In FIG. 9, when the number of transmitting antennas is M, the number of receiving antennas is N, and the number of rows selected from the channel matrix is S, the ratio of communication capacities is represented by six combinations of M, N, and S. For example, (3×3×2) indicates that the number of transmitting antennas is 3, the number of receiving antennas is 3, and the number of rows selected from the channel matrix is 2.
The communication capacity C is calculated by the above-mentioned Expression 11. In addition, it is assumed that each element of the channel matrix H has a complex normal distribution with a mean of 0 and a variance of 1.
As can be seen from FIG. 9, the communication capacity in each case is 90% to 95% of the communication capacity in the eigenmode transmission.
Next, the operation of the selecting circuit 212 selecting S rows from the channel matrix H by the patterns 1 to 5 described in the first embodiment of the present invention in order to improve the communication capacity will be described.
FIGS. 10A to 10F are bar graphs illustrating the ratio of communication capacities when the sub-channel matrices HS selected by the patterns 1 to 5 are used to calculate the antenna weighting coefficient matrix W. Similar to FIG. 9, FIGS. 10A to 10F plot the average value of the ratios of the communication capacity C1 when S data streams are transmitted by using the antenna weighting coefficient matrix W to the communication capacity C0 when S data streams are transmitted by eigenmode transmission (C1/C0) for 10000 samples.
In the bar graphs shown in FIGS. 10A to 10F, “sel0” in the horizontal axis indicates characteristics when S rows are appropriately selected from N rows of the channel matrix H of N rows and M columns to generate the sub-channel matrix HS, which are the same as those in FIG. 4. In addition, “sel0” to “sel5” in the horizontal axis indicate characteristics when the sub-channel matrices HS selected by the patterns 1 to 5 are generated, respectively.
FIG. 10A is a bar graph illustrating characteristics when (M×N×S) is (3×3×2), FIG. 10B is a bar graph illustrating characteristics when (M×N×S) is (4×4×2), and FIG. 10C is a bar graph illustrating characteristics when (M×N×S) is (4×4×3). In addition, FIG. 10D is a bar graph illustrating characteristics when (M×N×S) is (5×5×2), FIG. 10E is a bar graph illustrating characteristics when (M×N×S) is (5×5×3), and FIG. 10F is a bar graph illustrating characteristics when (M×N×S) is (5×5×4).
As can be seen from six bar graphs shown in FIGS. 10A to 10F, in all the patterns 1 to 5, communication characteristics are improved as compared to the case in which S rows are appropriately selected from N rows of the channel matrix H of N rows and M columns to generate the sub-channel matrix HS.
The method of calculating the antenna weighting coefficient matrix W according to the second embodiment of the present invention has been described with reference to FIG. 8. As described above, according to the second embodiment of the present invention, when the maximum value of the number of data streams to be subjected to a transmission beam-forming process is smaller than the number of antennas, the antenna weighting coefficient matrix W is calculated based on the channel matrix H and the sub-channel matrix HS obtained by extracting the number of rows corresponding to the maximum value of the number of data streams from the channel matrix H.
As such, since the antenna weighting coefficient matrix W is calculated based on the channel matrix H and the sub-channel matrix HS, it is possible to prevent an increase in the amount of calculation while maintaining characteristics at a certain level. When this wireless communication system is actually mounted to a communication apparatus, the amount of computation depends on the size of a circuit and a computation time. Therefore, it is possible to reduce the size of a circuit and shorten the computation time.
When the sub-channel matrix HS is generated, it is possible to improve characteristics while minimizing an increase in the amount of computation by giving row selection conditions, similar to the first embodiment.
Third Embodiment
In the second embodiment, the channel matrix H and the sub-channel matrix HS obtained by extracting the number of rows corresponding to the maximum value of the number of data streams from the channel matrix H are input to the arithmetic circuit 214 to calculate the antenna weighting coefficient matrix W. A third embodiment of the present invention is a combination of the first embodiment and the second embodiment of the present invention, and uses the channel matrix H and the sub-channel matrix HS to calculate the antenna weighting coefficient matrix W, which will be described below.
FIG. 11 is a diagram illustrating the structure of an arithmetic circuit 414 according to the third embodiment of the present invention. Hereinafter, the structure of the arithmetic circuit 414 according to the third embodiment of the present invention will be described with reference to FIG. 11.
The arithmetic circuit 414 shown in FIG. 11 replaces the arithmetic circuit 414 according to the first embodiment of the present invention shown in FIG. 2. That is, the arithmetic circuit 414 receives the sub-channel matrix HS selected by the selecting circuit 212 and the channel matrix H estimated by the channel estimating circuit 102, and outputs the antenna weighting coefficient matrix W for a transmission beam-forming.
As shown in FIG. 11, the arithmetic circuit 414 according to the third embodiment of the present invention includes first coefficient calculating circuits 422a, 422b, and 422c, matrix multiplying circuits 424a and 424b, and transposition circuits 426a, 426b, and 426c. 
The first coefficient calculating circuit 422a receives the channel matrix H and the sub-channel matrix HS, and calculates the temporary antenna weighting coefficient matrix WT based on the channel matrix H and the sub-channel matrix HS. The structure of the first coefficient calculating circuit 422a will be described below. The first coefficient calculating circuit 422a outputs the calculated temporary antenna weighting coefficient matrix WT to the matrix multiplying circuit 424a. 
The matrix multiplying circuit 424a is for multiplying matrices. The matrix multiplying circuit 424a according to this embodiment multiplies the channel matrix H estimated by the channel estimating circuit 102 by the temporary antenna weighting coefficient matrix WT calculated by the first coefficient calculating circuit 422a to generate a weighted channel matrix HW (HW=HWT). In this case, HW indicates a matrix of M rows and S columns. The weighted channel matrix HW generated by the matrix multiplying circuit 424a is output to the transposition circuit 426a. 
The transposition circuit 426a is for transposing a matrix. The transposition circuit 426a according to this embodiment receives the weighted channel matrix HW generated by the matrix multiplying circuit 424a, transposes the weighted channel matrix HW, and outputs a transposed weighted channel matrix HWT. The transposed weighted channel matrix HWT is input to the first coefficient calculating circuit 422b. 
The transposition circuit 426b is for transposing a matrix. The transposition circuit 426b according to this embodiment receives the channel matrix H estimated by the channel estimating circuit 102, transposes the channel matrix H, and outputs a transposed channel matrix HT. The transposed channel matrix HT is input to the first coefficient calculating circuit 422b and the matrix multiplying circuit 424b. 
The first coefficient calculating circuit 422b calculates a reverse antenna weighting coefficient matrix WR based on the transposed weighted channel matrix HWT output from the transposition circuit 426a and the transposed channel matrix HT output from the transposition circuit 426b. The first coefficient calculating circuit 422b outputs the calculated reverse antenna weighting coefficient matrix WR to the matrix multiplying circuit 424b. 
The matrix multiplying circuit 424b is for multiplying matrices. The matrix multiplying circuit 424b according to this embodiment multiplies the reverse antenna weighting coefficient matrix WR output from the first coefficient calculating circuit 422b by the transposed channel matrix HT output from the transposition circuit 426b to generate a reverse weighted channel matrix HWR (HWR=HTWR). In the case, the reverse weighted channel matrix HWR includes N rows and S columns. The reverse weighted channel matrix HWR generated by the matrix multiplying circuit 424b is input to the transposition circuit 426c. 
The transposition circuit 426c is for transposing a matrix. The transposition circuit 426c according to this embodiment transposes the reverse weighted channel matrix HWR generated by the matrix multiplying circuit 424b to generate a transposed reverse weighted channel matrix HWRT. The transposed reverse weighted channel matrix HWRT is input to the first coefficient calculating circuit 422c. 
The first coefficient calculating circuit 422c receives the channel matrix H estimated by the channel estimating circuit 102 and the transposed reverse weighted channel matrix HWRT output from the transposition circuit 426c, and generates the antenna weighting coefficient matrix W based on the channel matrix H and the transposed reverse weighted channel matrix HWRT. The first coefficient calculating circuit 422c outputs the calculated antenna weighting coefficient matrix W to the weighting circuit 110.
The antenna weighting coefficient matrix W output from the first coefficient calculating circuit 422c may be input to the matrix multiplying circuit 424a again. It is possible to improve communication characteristics by repeatedly performing a computing process using the antenna weighting coefficient matrix W output from the first coefficient calculating circuit 422c. 
The structure of the arithmetic circuit 414 according to the third embodiment of the present invention has been described above with reference to FIG. 11. Next, the structure of a first coefficient calculating circuit according to the third embodiment of the present invention will be described.
FIG. 12 is a diagram illustrating the structure of the first coefficient calculating circuit 422a according to the third embodiment of the present invention. Hereinafter, the structure of the first coefficient calculating circuit 422a according to the fourth embodiment of the present invention will be described with reference to FIG. 12.
The first coefficient calculating circuit 422a receives a sub-channel matrix A and a channel matrix B, and calculates an antenna weighting coefficient matrix WA. In the arithmetic circuit 414 shown in FIG. 11, three first coefficient calculating circuits are used. The relationship among A, B, and WA in the first coefficient calculating circuit 422a is (HS, H, WT), the relationship among A, B, and WA in the first coefficient calculating circuit 422b is (HWT, Hr, WR), and the relationship among A, B, and WA in the first coefficient calculating circuit 422c is (HWRT, H, W). In this embodiment, only the first coefficient calculating circuit 422a is described, but the first coefficient calculating circuits 422b and 422c have the same structure as the first coefficient calculating circuit 422a. 
As shown in FIG. 12, the first coefficient calculating circuit 422a according to the third embodiment of the present invention includes a second coefficient calculating circuit 432, matrix multiplying circuits 434a and 434b, and a SVD circuit 436. The structure of the first coefficient calculating circuit 422a shown in FIG. 11 is the same as that of the arithmetic circuit 214 according to the second embodiment of the present invention shown in FIG. 7.
The second coefficient calculating circuit 432 calculates a temporary antenna weighting coefficient matrix WAT based on the sub-channel matrix A (in this embodiment, the sub-channel matrix HS). For example, the second coefficient calculating circuit 432 can use singular value decomposition to calculate the temporary antenna weighting coefficient matrix WAT. When singular value decomposition is performed on the sub-channel matrix A, the following Expression 15 is obtained.

A=UDVH  (Expression 15)

When the temporary antenna weighting coefficient matrix WAT is calculated by singular value decomposition, the second coefficient calculating circuit 432 uses the matrix (right singular matrix) V of Expression 15 as the temporary antenna weighting coefficient matrix WAT. The second coefficient calculating circuit 432 outputs the calculated temporary antenna weighting coefficient matrix WAT to the matrix multiplying circuits 434a and 434b. 
The matrix multiplying circuit 434a is for multiplying matrices. The matrix multiplying circuit 434a according to this embodiment multiplies the channel matrix B (in this embodiment, the channel matrix H) by the temporary antenna weighting coefficient matrix WAT calculated by the second coefficient calculating circuit 432 to generate a weighted channel matrix HAW (HAW=BWAT). In this case, HAW indicates a matrix of M rows and S columns. The weighted channel matrix HAW generated by the matrix multiplying circuit 434a is output to the SVD circuit 436.
The SVD circuit 436 is for performing singular value decomposition. In this embodiment, the SVD circuit 436 receives the weighted channel matrix HAW generated by the matrix multiplying circuit 434a and performs the singular value decomposition on the weighted channel matrix HAW. When the singular value decomposition is performed on the weighted channel matrix HAW, the following Expression 16 is obtained.

HAW=UDVH  (Expression 16)

The matrix (right singular matrix) V obtained by the Expression 16 is output to the matrix multiplying circuit 434b. 
The matrix multiplying circuit 434b is for multiplying matrices. The matrix multiplying circuit 434b according to this embodiment multiplies the temporary antenna weighting coefficient matrix WAT calculated by the second coefficient calculating circuit 432 by the matrix V calculated by the SVD circuit 436 to generate the antenna weighting coefficient matrix WA (WA=WATV). In this embodiment, the antenna weighting coefficient matrix WA is the temporary antenna weighting coefficient matrix WT. The rows that are not used in the calculation of the temporary antenna weighting coefficient matrix WAT (that is, the rows of the channel matrix B that are not included in the sub-channel matrix A) are used to calculate the antenna weighting coefficient matrix WA. Therefore, it is possible to improve characteristics.
The antenna weighting coefficient matrix WA generated by the matrix multiplying circuit 434b is the reverse antenna weighting coefficient matrix WR in the first coefficient calculating circuit 422b, and is the antenna weighting coefficient matrix W in the first coefficient calculating circuit 422c. 
The structure of the first coefficient calculating circuit 422a according to the third embodiment of the present invention has been described with reference to FIG. 12. Next, a method of calculating the antenna weighting coefficient matrix W according to the third embodiment of the present invention will be described.
FIG. 13 is a flowchart illustrating the method of calculating the antenna weighting coefficient matrix W according to the third embodiment of the present invention. Hereinafter, the method of calculating the antenna weighting coefficient matrix W according to the third embodiment of the present invention will be described with reference to FIG. 13.
When the channel matrix H of N rows and M columns is input to the weighting coefficient matrix calculating circuit 104, the selecting circuit 112 appropriately selects S rows from the channel matrix H and generates the sub-channel matrix HS (Step S402). When the selecting circuit 112 generates the sub-channel matrix HS, the first coefficient calculating circuit 422a calculates the temporary antenna weighting coefficient matrix WT based on the generated sub-channel matrix HS and the channel matrix H (Step S404). As described above, for example, the first coefficient calculating circuit 422a can use singular value decomposition to calculate the temporary antenna weighting coefficient matrix WT.
When the temporary antenna weighting coefficient matrix WT is completely generated in Step S404, the matrix multiplying circuit 424a multiplies the channel matrix H by the temporary antenna weighting coefficient matrix WT to generate the weighted channel matrix HW (Step S406).
When the weighted channel matrix HW is generated in Step S406, the transposition circuit 426a transposes the weighted channel matrix HW. Then, the first coefficient calculating circuit 422b receives the transposed weighted channel matrix HWT and the transposed matrix HT of the channel matrix H, and calculates the reverse antenna weighting coefficient matrix WR using the transposed weighted channel matrix HWT and the transposed matrix HT of the channel matrix H (Step S408).
When the reverse antenna weighting coefficient matrix WR is generated in Step S408, the matrix multiplying circuit 424b multiplies the transposed channel matrix HT transposed from the channel matrix H by the transposition circuit 426b by the reverse antenna weighting coefficient matrix WR. A reverse weighted channel matrix HWR is obtained by the multiplication of the transposed channel matrix HT and the reverse antenna weighting coefficient matrix WR (Step S410).
When the reverse weighted channel matrix HWR is obtained in Step S410, the transposition circuit 426c transposes the reverse weighted channel matrix HWR to obtain a transposed reverse weighted channel matrix HWRT. Then, the first coefficient calculating circuit 422c receives the transposed reverse weighted channel matrix HWRT and the channel matrix H and calculates the antenna weighting coefficient matrix W (Step 412).
In this way, finally, the matrix is transposed to obtain the antenna weighting coefficient matrix W. Therefore, it is possible to obtain the same effects as those when the arithmetic circuit 414 performs a two-way beam-forming process.
From the relationship of S<min(M, N), since the rank of the matrix subjected to singular value decomposition is less than that in eigenmode transmission, the amount of calculation is reduced by a value corresponding to the reduction in the rank of the matrix. In addition, the amount of calculation is reduced by a value corresponding to a reduction in the number of elements of the matrix generated by the singular value decomposition. In addition, in order to improve communication characteristics, Step S406 to Step S412 may be repeatedly performed using the antenna weighting coefficient matrix W output from the first coefficient calculating circuit 422c. As described above, the total number of processes is increased, but the amount of calculation of the complicated singular value decomposition process is reduced. Therefore, it is possible to prevent an increase in the total amount of computation.
The first coefficient calculating circuits 422a, 422b, and 422c may be integrated with each other, the matrix multiplying circuits 424a, 424b, 424c, 434a, and 434b may be integrated with each other, and the transposition circuits 426a, 426b, and 426c may be integrated with each other according to conditions. In addition, the second coefficient calculating circuit 432 and the SVD circuit 436 may be integrated with each other.
FIGS. 14A to 14F are line graphs illustrating the comparison between eigenmode transmission and transmission using the antenna weighting coefficient matrix W calculated by the method of calculating the antenna weighting coefficient matrix W according to the third embodiment of the present invention. FIGS. 14A to 14F plot the maximum value and the minimum value of the ratios of communication capacity C1 when S data streams are transmitted by the antenna weighting coefficient matrix W to communication capacity C0 when S data streams are transmitted by eigenmode transmission (C1/C0) for 10000 samples. In the line graphs, the horizontal axis indicates the number of times Steps S406 to S412 are repeated. In addition, the number of repetitions is zero when the temporary antenna weighting matrix WT calculated by the first coefficient calculating circuit 422a is used instead of the antenna weighting coefficient matrix W.
In FIGS. 14A to 14F, when the number of transmitting antennas is M, the number of receiving antennas is N, and the number of rows selected from the channel matrix is S, the ratio of communication capacities is represented by six combinations of M, N, and S.
Here, the communication capacity C is calculated by the above-mentioned Expression 9. In addition, it is assumed that each element of the channel matrix H has a complex normal distribution with a mean of 0 and a variance of 1.
FIG. 14A is a line graph illustrating characteristics when (M×N×S) is (3×3×2), FIG. 14B is a line graph illustrating characteristics when (M×N×S) is (4×4×2), and FIG. 14C is a line graph illustrating characteristics when (M×N×S) is (4×4×3). In addition, FIG. 14D is a line graph illustrating characteristics when (M×N×S) is (5×5×2), FIG. 14E is a line graph illustrating characteristics when (M×N×S) is (5×5×3), and FIG. 14F is a line graph illustrating characteristics when (M×N×S) is (5×5×4).
As can be seen from six line graphs shown in FIGS. 14A to 14F, similar to the line graphs according to the first embodiment of the present invention, in all the cases, the ratio of the communication capacities converges on 1. That is, the communication capacity when the antenna weighting coefficient matrix W is used converges on the communication capacity during the eigenmode transmission.
The method of calculating the antenna weighting coefficient matrix W according to the third embodiment of the present invention has been described above. Next, modifications of the third embodiment of the present invention will be described.
FIG. 15 is a diagram illustrating the structure of an arithmetic circuit 414′ according to a modification of the third embodiment of the present invention. Hereinafter, the structure of the arithmetic circuit 414′ according to the modification of the third embodiment of the present invention will be described with reference to FIG. 15.
The arithmetic circuit 414′ shown in FIG. 15 includes second coefficient calculating circuits 432a and 432b, instead of the first coefficient calculating circuits 422b and 422c of the arithmetic circuit 414 shown in FIG. 11. Even when the arithmetic circuit 414′ has the above-mentioned structure, it is possible to obtain the same communication characteristics as those obtained by the arithmetic circuit 414 shown in FIG. 11.
Combinations of the first coefficient calculating circuit and the second coefficient calculating circuit are not limited to the above. For example, the second coefficient calculating circuit may be provided instead of the first coefficient calculating circuit 422b, and the second coefficient calculating circuit may be provided instead of the first coefficient calculating circuit 422c. 
The structure of the arithmetic circuit 414′ according to the modification of the third embodiment of the present invention has been described above. Next, the operation of the selecting circuit 112 selecting S rows from the channel matrix H by the patterns 1 to 5 described in the first embodiment of the present invention in order to improve the convergence speed of the communication capacity which will be described.
FIGS. 16A to 16F are line graphs illustrating the convergence of communication capacity when the antenna weighting coefficient matrix W is calculated using the sub-channel matrix HS selected by the patterns 1 to 5. In FIGS. 16A to 16F show the antenna weighting coefficient matrix W calculated using the sub-channel matrix HS that is selected by the patterns 1 to 5 when (M×N×S) is (3×3×2).
FIG. 16A is the same as FIG. 14A, FIG. 16B is a line graph illustrating characteristics when the sub-channel matrix HS selected by the pattern 1 is used, and FIG. 16C is a line graph illustrating characteristics when the sub-channel matrix HS selected by the pattern 2 is used. FIG. 16D is a line graph illustrating characteristics when the sub-channel matrix HS selected by the pattern 3 is used, FIG. 16E is a line graph illustrating characteristics when the sub-channel matrix HS selected by the pattern 4 is used, and FIG. 16F is a line graph illustrating characteristics when the sub-channel matrix HS selected by the pattern 5 is used.
As can be seen from the graphs shown in FIGS. 16A to 16F, similar to the line graphs shown in the first embodiment of the present invention, in all the patterns 1 to 5, convergence characteristics are improved, as compared to when S rows are appropriately selected from N rows of the channel matrix H of N rows and M columns to generate the sub-channel matrix HS. The improvement in the convergence characteristics makes it possible to shorten a calculation time.
In this embodiment, (M×N×S) is (3×3×2), but other combinations of M, N, and S may be used. In this case, the convergence characteristics are also improved, as compared to when S rows are appropriately selected from N rows of the channel matrix H of N rows and M columns to generate the sub-channel matrix HS.
As described above, according to the third embodiment of the present invention, when the maximum value of the number of data streams to be subjected to a transmission beam-forming process is smaller than the number of antennas, the antenna weighting coefficient matrix W is calculated based on the channel matrix H and the sub-channel matrix HS obtained by extracting the number of rows corresponding to the maximum value of the number of data streams from the channel matrix H.
As such, since the antenna weighting coefficient matrix W is calculated based on the channel matrix H and the sub-channel matrix HS, the rank of the matrix used for calculation is reduced, and it is possible to reduce the amount of calculation of SVD, which is a representative example of the matrix computation, while maintaining characteristics at a certain level. In addition, it is possible to prevent deterioration of characteristics and reduce the overall size of a weighting coefficient matrix calculating circuit. When this wireless communication system is actually mounted to a communication apparatus, the amount of computation depends on the size of a circuit and a computation time. Therefore, it is possible to reduce the size of a circuit and shorten the computation time.
When the sub-channel matrix HS is generated, it is possible to improve convergence characteristics while minimizing an increase in the amount of computation by giving row selection conditions, similar to the first and second embodiments of the present invention.
In addition, a computer program for estimating a matrix by the above-mentioned method or performing an operation on a matrix may be stored in the transmitter 100 or the receiver 200, and a computing device, such as a CPU (central processing unit) may read the stored computer programs and execute it to implement the method of calculating the antenna weighting coefficient matrix W according to any one of the first to third embodiments of the present invention.
The present application contains subject matter related to that disclosed in Japanese Priority Patent Application JP 2008-120598 filed in the Japan Patent Office on May 2, 2008, the entire content of which is hereby incorporated by reference.
It should be understood by those skilled in the art that various modifications, combinations, sub-combinations and alterations may occur depending on design requirements and other factors insofar as they are within the scope of the appended claims or the equivalents thereof.

## Claims

1. A wireless communication apparatus comprising:
a matrix estimating unit that estimates a channel matrix of N rows and M columns (N and M are natural numbers);
a selecting unit that selects S rows (S is a natural number, and S<MIN(M, N)) from the channel matrix estimated by the matrix estimating unit and generates a sub-channel matrix of S rows and M columns; and
an arithmetic unit that calculates an antenna weighting coefficient matrix based on the channel matrix estimated by the matrix estimating unit and the sub-channel matrix generated by the selecting unit,
wherein the arithmetic unit includes:
a weighted channel matrix generating unit that multiplies the channel matrix by a temporary antenna weighting coefficient matrix calculated based on the sub-channel matrix to generate a weighted channel matrix;
a reverse weighted channel matrix generating unit that transposes the channel matrix and the weighted channel matrix, and multiplies the transposed channel matrix by a reverse antenna weighting coefficient matrix calculated based on the transposed weighted channel matrix to generate a reverse weighted channel matrix; and
an antenna weighting coefficient matrix calculating unit that calculates the antenna weighting coefficient matrix based on a transposed reverse weighted channel matrix.

2. The wireless communication apparatus according to claim 1,
wherein the weighted channel matrix generating unit repeatedly generates the weighted channel matrix using the antenna weighting coefficient matrix calculated by the antenna weighting coefficient matrix calculating unit instead of the temporary antenna weighting coefficient matrix.

3. The wireless communication apparatus according to claim 1,
wherein the arithmetic unit includes:
a first coefficient calculating unit that calculates the temporary antenna weighting coefficient matrix based on the sub-channel matrix;
a first matrix multiplying unit that multiplies the channel matrix by the temporary antenna weighting coefficient matrix to generate the weighted channel matrix;
a first transposition unit that transposes the channel matrix and the weighted channel matrix;
a second coefficient calculating unit that calculates the reverse antenna weighting coefficient matrix based on the transposed weighted channel matrix;
a second matrix multiplying unit that multiplies the transposed channel matrix by the reverse antenna weighting coefficient matrix to generate the reverse weighted channel matrix;
a second transposition unit that transposes the reverse weighted channel matrix; and
a third coefficient calculating unit that calculates the antenna weighting coefficient matrix based on the transposed reverse weighted channel matrix.

4. The wireless communication apparatus according to claim 3,
wherein the first coefficient calculating unit calculates the temporary antenna weighting coefficient matrix based on the sub-channel matrix and the channel matrix,
the second coefficient calculating unit calculates the reverse antenna weighting coefficient matrix based on the transposed weighted channel matrix and the transposed channel matrix, and
the third coefficient calculating unit calculates the antenna weighting coefficient matrix based on the transposed reverse weighted channel matrix and the channel matrix.

5. The wireless communication apparatus according to claim 3,
wherein at least one of the first to third coefficient calculating units includes:
a fourth coefficient calculating unit that calculates the temporary antenna weighting coefficient matrix based on the sub-channel matrix;
a third matrix multiplying unit that multiplies the channel matrix by the temporary antenna weighting coefficient matrix to generate the weighted channel matrix;
a singular value decomposing unit that performs singular value decomposition on the weighted channel matrix; and
a fourth matrix multiplying unit that multiplies a matrix obtained by the singular value decomposition by the temporary antenna weighting coefficient matrix to generate the antenna weighting coefficient matrix.

6. The wireless communication apparatus according to claim 1,
wherein the selecting unit selects the S rows that allow the sum of the squares of elements of a matrix to be the maximum.

7. The wireless communication apparatus according to claim 1,
wherein the selecting unit selects the S rows that allow an eigenvalue of a covariance matrix of a matrix or the maximum value of a singular value of the matrix to be the maximum.

8. The wireless communication apparatus according to claim 1,
wherein the selecting unit selects the S rows that allow an eigenvalue of a covariance matrix of a matrix or the minimum value of a singular value of the matrix to be the maximum.

9. The wireless communication apparatus according to claim 1,
wherein the selecting unit selects the S rows that allow a determinant of a covariance matrix of a matrix to be the maximum.

10. The wireless communication apparatus according to claim 1,
wherein the selecting unit selects the S rows that allow a value obtained by dividing a determinant of a covariance matrix of a matrix by the sum of the squares of elements of the matrix to be the maximum.

11. A wireless communication method comprising the steps of:
estimating a channel matrix of N rows and M columns (N and M are natural numbers);
selecting S rows (S is a natural number, and S<MIN(M, N)) from the estimated channel matrix and generating a sub-channel matrix of S rows and M columns; and
calculating an antenna weighting coefficient matrix based on the estimated channel matrix and the generated sub-channel matrix,
wherein the calculating step includes the sub-steps of:
multiplying the channel matrix by a temporary antenna weighting coefficient matrix calculated based on the sub-channel matrix to generate a weightec channel matrix;
transposing the channel matrix and the weighted channel matrix, and multiplying the transposed channel matrix by a reverse antenna weighting coefficien matrix calculated based on the transposed weighted channel matrix to generate a reverse weighted channel matrix; and
calculating the antenna weighting coefficient matrix based on a transposed reverse weighted channel matrix.

12. A wireless communication system comprising:
a first wireless communication apparatus; and
a second wireless communication apparatus,
wherein the first wireless communication apparatus includes a transmitting unit that transmits reference signals to the second wireless communication apparatus through a plurality of antennas,
the second wireless communication apparatus includes:
a matrix estimating unit that estimates a channel matrix of N rows and M columns (N and M are natural numbers) based on the reference signals received through a plurality of antennas;
a selecting unit that selects S rows (S is a natural number, and S<MIN(M, N)) from the channel matrix estimated by the matrix estimating unit and generates a sub-channel matrix of S rows and M columns; and
an arithmetic unit that calculates an antenna weighting coefficient matrix based on the channel matrix estimated by the matrix estimating unit and the sub-channel matrix generated by the selecting unit, and
the arithmetic unit includes:
a weighted channel matrix generating unit that multiplies the channel matrix by a temporary antenna weighting coefficient matrix calculated based on the sub-channel matrix to generate a weighted channel matrix;
a reverse weighted channel matrix generating unit that transposes the channel matrix and the weighted channel matrix, and multiplies the transposed channel matrix by a reverse antenna weighting coefficient matrix calculated based on the transposed weighted channel matrix to generate a reverse weighted channel matrix; and
an antenna weighting coefficient matrix calculating unit that calculates the antenna weighting coefficient matrix based on a transposed reverse weighted channel matrix.