# Patent

## Title
Secondary side post regulators

## Abstract

A synchronously switched buck post regulator is revealed for multi-output forward converters. The synchronously switched buck post regulator accomplishes precise independent load regulation for each output and reduced magnetics volume by using a coupled inductor with a common core for all outputs plus a second smaller inductor for each output except the highest voltage output.

## Background
CROSS-REFERENCE TO RELATED APPLICATION
This is a division of application Ser. No. 10/944,588, filed 2004 Sep. 18, now abandoned.


BACKGROUND OF THE INVENTION
1. Field of the Invention
The subject invention generally pertains to electronic power conversion circuits, and more specifically to high frequency, switched mode electronic power converters. Some of the subject matter of this application was first revealed in Disclosure Document Number 527396.
2. Description of Related Art
Many electronic circuits require power at more than one voltage, so power supply manufacturers offer power supplies with multiple regulated outputs. This was often an application for magnetic amplifiers, but magnetic amplifiers have become less popular as switching frequencies have risen and silicon semiconductor prices have fallen. One common way of providing additional regulated outputs in current use is illustrated in FIG. 1. Operational wave forms for the FIG. 1 circuit are illustrated in FIG. 2. In FIG. 1 a second regulated output is provided by a synchronously switched buck post regulator, which is a sort of semiconductor magnetic amplifier. In the FIG. 1 circuit an output choke L1 is provided for the main output and a second output choke L2 is provided for the second output. In general the first output voltage is fed back to control the primary switch and the second output is regulated with a local feedback loop which controls the delay in the turn on time of MS2A with respect to the turn on time of MS1A. This is leading edge modulation which is similar to that used in a magnetic amplifier. In most of these power converters there is a maximum power level which is the sum of the powers of the two outputs, but either of the two outputs can be operated at or near the maximum power level. This requires two output chokes each of which must be sized for the converter's maximum power level. An alternate method is illustrated in FIG. 3. The FIG. 3 circuit uses a coupled output choke to obviate two separate magnetic circuit elements sized for maximum power. In the coupled inductor the single core must be sized for maximum power and the window area must be sized to accommodate two windings each of which must be capable of accommodating the maximum output power. This is an improvement which results in a smaller and cheaper output choke, compared to the FIG. 1 circuit. The disadvantage of the FIG. 3 circuit is that the second output is not independently regulated and two secondary windings are required on the main transformer. What is needed is a circuit that achieves the performance of the FIG. 1 circuit but with lower total output choke volume.
OBJECTS AND ADVANTAGES
An object of the subject invention is to reveal a technique for achieving multiple independently regulated output voltages with reduced total magnetics volume.
Further objects and advantages of my invention will become apparent from a consideration of the drawings and ensuing description.
These and other objects of the invention are provided by new circuit structures for achieving independently regulated outputs in a multi-output power supply with reduced magnetics volume. Also revealed is a simple capacitor coupled floating drive circuit that is tolerant of voltage variations at the terminals of the high side switch.



BRIEF DESCRIPTION OF THE DRAWINGS
The present invention is illustrated by reference to the drawings.
FIG. 1 illustrates a forward converter with two independently regulated outputs according to the prior art.
FIG. 2(a) illustrates the timing wave form for the MPRI and MS1A switches of FIG. 1.
FIG. 2(b) illustrates the timing wave form for the MS1B switch of FIG. 1.
FIG. 2(c) illustrates the timing wave form for the MS2A switch of FIG. 1.
FIG. 2(d) illustrates the timing wave form for the MS2B switch of FIG. 1.
FIG. 2(e) illustrates the current wave form for the L1 inductor of FIG. 1.
FIG. 2(f) illustrates the current wave form for the L2 inductor of FIG. 1.
FIG. 3 illustrates a forward converter with two outputs utilizing a single coupled output choke according to the prior art.
FIG. 4 illustrates a forward converter with two independently regulated outputs using a coupled output choke according to the subject invention.
FIG. 5(a) illustrates the timing wave form for the MPRI and MS1A switches of FIG. 4 according to the subject invention.
FIG. 5(b) illustrates the timing wave form for the MS1B switch of FIG. 4 according to the subject invention.
FIG. 5(c) illustrates the timing wave form for the MS2A switch of FIG. 4 according to the subject invention.
FIG. 5(d) illustrates the timing wave form for the MS2B switch of FIG. 4 according to the subject invention.
FIG. 5(e) illustrates the current wave form for the L1A inductor of FIG. 4 according to the subject invention.
FIG. 5(f) illustrates the current wave form for the L2 and L1B inductors of FIG. 4 according to the subject invention.
FIG. 6 illustrates a forward converter with two independently regulated outputs using a coupled output choke with low magnetic coupling according to the subject invention.
FIG. 7 illustrates a current doubler rectifier forward converter with a quarter bridge primary network and two independently regulated outputs using coupled output chokes according to the subject invention.



SUMMARY
The subject invention reveals a circuit that can achieve multiple independently regulated outputs in a forward converter with lower overall magnetic circuit volume and cost.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
FIG. 1 illustrates a single-ended forward converter with two outputs that can be independently regulated. The first output relies on modulation of the primary switch to achieve its regulation. The second output relies upon a second local control loop (not shown) to accomplish its regulation. The second output is a synchronously switched buck converter which is also referred to as a semiconductor magnetic amplifier since the switch control is similar to that of a magnetic amplifier. The main switch, MS2A, of the second output's buck converter is turned off in synchronization with the first output in each switching cycle and turned on after the switch MS1A during each switching cycle. Wave forms of the switch timing and output choke currents are illustrated in FIGS. 2(a) through 2(f). In commercial multi-output power supplies the designer's goal is to provide two independently regulated outputs, each of which can handle the converter's full output power rating in order to accommodate the broadest range of customers. In the FIG. 1 circuit each of the two output chokes must be sized to accommodate the full output load. FIG. 3 illustrates another prior art example that has been used for multi-output power supplies. In the FIG. 3 circuit there is a single output choke that accommodates both outputs, but the two outputs cannot be independently regulated in the FIG. 3 circuit. Different voltage levels can be set by adjusting the turns ratios for the two outputs. The turns ratio for the secondary windings should match the turns ratio for the coupled inductor windings in order for the output currents to be non-pulsating. The FIG. 3 circuit has the advantage of smaller magnetics over the FIG. 2 circuit, since the two outputs share a common magnetic core, but the two outputs are not independently regulated. The common magnetic core of the two outputs of the FIG. 3 circuit need only be large enough to handle the full power of the power supply.
FIG. 4 illustrates a two output power supply with independently regulated outputs that uses a common magnetic core for the two outputs, thereby saving the space of another large core sized to handle the full output power, plus a second much smaller core, L2, with a much smaller inductance that adds the ability to independently regulate the second output in a synchronously switched buck converter. In the FIG. 4 circuit the first output is controlled by controlling the duty cycle of the primary switch MPRI. The switch MS1A is operated in full synchronization with MPRI and the switch MS1B is operated in anti-synchronization to MPRI. A wave form illustrating the switch timing for MPRI and MS1A is illustrated in FIG. 5(a). A wave form illustrating the timing for the MS1B switch is illustrated in FIG. 5(b). The output voltage for the first output is determined by the duty cycle of MPRI and the turns ratio of the transformer T. For the second output the switch MS2A is turned off in synchronization with MPRI, but MS2A is turned on after MPRI so that there is a delay between the time MPRI is turned on and the time at which MS2A is turned on, as illustrated in FIG. 5(c). Switch MS2B turns on in synchronization with switch MS1B when MPRI is turned off, but MS2B turns off after MS1B is turned off due to the delay in the turn on of MS2A. MS2B operates in anti-synchronization to MS2A, as illustrated in FIG. 5(d). Because MS2A is delayed with respect to MPRI and MS1A the duty cycle for the second output will be less than the duty cycle for the first output and the voltage at the second output will be less than the voltage at the first output. The inductor L2 forces the output current for the second output to be non-pulsating since an infinite voltage would be required to create a step change in current in L2. The current in L1A is also non-pulsating, since there is no mechanism in the circuit to instantly interrupt current flow in either L1A or L1B. Current in L1A is provided through MS1A and the transformer secondary winding when MPRI is on or through MS1B when MPRI is off, although L1A could still be pulsating despite this fact, but only if the current in L1B is interrupted. L2 forces the current in L1B to be continuous. The currents in L1A and L1B will both be continuous, but they will not, in general, be simple triangular wave forms. A current wave form for L1A is illustrated in FIG. 5(e) and a corresponding wave form for L1B current is illustrated in FIG. 5(f). The current in L1B must be equal to the current in L2, since the two windings are in series. The slope of the current in L2 will depend on the value of L2 and the voltage applied to L2. During the course of one switching cycle there are three different values of applied voltage for L2. When MPRI is off MS1B and MS2B are on and the voltage at the node A is zero. While MPRI is off the dotted terminal of L2 is at ground and the undotted terminal of L2 is at a voltage V1, where




  
    
      V
      1
    
    =
    
      
        V
        OUT2
      
      -
      
        
          
            N
            2
          
          
            N
            1
          
        
        *
        
          V
          OUT1
        
      
    
  
  ,




VOUT2 is the output voltage of the second output, VOUT1 is the output voltage of the first output, N2 is the number of turns of L1B, and N1 is the number of turns of L1A. Depending on the turns ratio,





  
    
      N
      2
    
    
      N
      1
    
  
  ,




the current slope of L2 during the time that MPRI is off can be positive, negative, or zero. In the example illustrated the current slope of L2 is near zero during the off time of MPRI, as illustrated in FIG. 5(f). When MPRI turns on, MS1A also turns on, and the voltage at the node A becomes positive, such that





  
    
      V
      A
    
    =
    
      
        
          N
          SEC
        
        
          N
          PRI
        
      
      *
      
        V
        INPUT
      
    
  
  ,




where VINPUT is the voltage of the input source, NPRI is the number of primary turns of transformer T, and NSEC is the number of secondary turns of transformer T. While MS1A is on and MS2B remains on, the voltage at the undotted terminal of L2 is V2, where





  
    
      V
      2
    
    =
    
      
        V
        OUT2
      
      +
      
        
          
            N
            2
          
          
            N
            1
          
        
        *
        
          (
          
            
              V
              A
            
            -
            
              V
              OUT1
            
          
          )
        
      
    
  
  ,




so that the undotted terminal of L2 is higher in voltage than the dotted terminal of L2 and the current in L2 must be decreasing. In the example illustrated the rate of decreasing current is relatively large for this example while MPRI is on and MS2A is off, as illustrated in FIG. 5(f). When MS2B turns off and MS2A turns on, while MPRI and MS1A remain on, the voltage at the dotted terminal of L2 is VA and the voltage at the undotted terminal of L2 is V2 and the voltage applied to L2 is V3, where V3=VA−V2. During the on time of MS2A, the current slope in L2 can be positive, negative, or zero depending on the turns ratio,





  
    
      N
      2
    
    
      N
      1
    
  
  ,




and the values of VA, VOUT1, and VOUT2. In the example illustrated the current slope in L2 is positive, as illustrated in FIG. 5(f). In a practical application the current slope of L2 during the on time of MS2A will, in general, be positive. Since L2 is connected in series with L1B the current in L1B will always be equal to the current in L2. The current slope in L1A will depend on the voltage applied to L1A and on the current slope of L1B and L2. The relationship between the voltage applied to L1A and its current slope is





  
    
      
        
          
            V
            L1A
          
          =
          
            
              
                
                  L
                  
                    1
                    ⁢
                    A
                  
                
                *
                
                  
                    dI
                    L1A
                  
                  dt
                
              
              +
              
                
                  M
                  12
                
                *
                
                  
                    dI
                    L1B
                  
                  dt
                
              
            
            =
            
              
                
                  L
                  
                    1
                    ⁢
                    A
                  
                
                *
                
                  
                    dI
                    L1A
                  
                  dt
                
              
              +
              
                
                  M
                  12
                
                *
                
                  
                    dI
                    L2
                  
                  dt
                
              
            
          
        
        ,
      
    
    
      
        (
        1
        )
      
    
  




where VL1A is the L1A applied voltage, IL1A is the current in L1A, IL1B is the L1B current, M12 is the L1AL1B mutual inductance, and IL2 is the L2 current. From equation (1) we can solve for the L1A current slope so that





  
    
      
        
          
            dI
            L1A
          
          dt
        
        =
        
          
            
              V
              L1A
            
            
              L
              
                1
                ⁢
                A
              
            
          
          -
          
            
              
                M
                12
              
              
                L
                
                  1
                  ⁢
                  A
                
              
            
            *
            
              
                
                  dI
                  L2
                
                dt
              
              .
            
          
        
      
    
    
      
        (
        2
        )
      
    
  




From equation (2) we see how the L1A current slope depends on both the voltage applied to the L1A winding and the current slope in the L1B winding. The turns ratio,





  
    
      N
      2
    
    
      N
      1
    
  
  ,




should be set to provide a minimum average applied voltage for L2 in order to minimize the current slopes and output ripple currents, which serves to minimize the size of L2. The primary benefit of the subject invention is a reduction in total magnetics volume combined with independent regulation of both outputs. The reduction of magnetics volume is accomplished by using a common core for the L1A and L1B windings so that only one large core capable of handling the full output power is required. A second smaller inductor, L2, is required to provide independent regulation for both outputs.

FIG. 5 illustrates the subject invention with series inductances l1A and l1B, which represent the leakage inductances or uncoupled inductances in each winding of L1. In general, it is preferable to loosely couple the windings in a coupled inductor forward converter application, thereby increasing the leakage inductances, since the main advantage of using the common core for both outputs is a size and cost reduction, not a performance improvement. The effect of the leakage inductance here is to improve the overall ripple performance. The relatively large leakage inductances, in comparison to a tightly coupled L1, provide additional current smoothing without an additional magnetic circuit element.
FIG. 7 illustrates a multi-output current doubler rectifier forward converter with a quarter bridge primary switching network and independently regulated outputs according to the subject invention. In the FIG. 7 circuit planar magnetics become more practical because of the reduction of primary winding voltage stress. In the current doubler configuration two chokes are required for each output. In the FIG. 7 circuit two independently regulated outputs are achieved using two large chokes and two small chokes.
CONCLUSION, RAMIFICATIONS, AND SCOPE OF INVENTION
Thus the reader will also see that a multi-output power supply using a secondary side synchronously switched post buck regulator can be improved by combining the output chokes onto a single core capable of handling the converter's full power and adding a small choke for the second output to provide fully independent regulation of the outputs.
While my above description contains many specificities, these should not be construed as limitations on the scope of the invention, but rather, as exemplifications or preferred embodiments thereof. Many other variations are possible. For example, power converters with more than two independently regulated outputs using a single large output coupled inductor plus much smaller chokes for the second and successive outputs are possible using the techniques revealed here.
Accordingly, the scope of the invention should be determined not by the embodiments illustrated, but by the appended claims and their legal equivalents.

## Claims

1. A secondary side buck post regulator for a second output of a multi-output forward converter comprising,
a first coupled inductor having a first winding and a second winding with an undotted terminal of said first winding connected to a first output of said multi-output forward converter and with said second winding connected in series with said second output of said multi-output forward converter,
a second inductor connected in series with said second winding of said first coupled inductor,
first switch means having a drain terminal connected to a dotted terminal of said first winding of said first coupled inductor,
second switch means, operable substantially in anti-synchronization to said first switch means, having a source terminal connected to an output terminal, which is common to said first output and to said second output of said multi-output forward converter and having a drain terminal connected to a source terminal of said first switch means and connected to a series winding network comprising said second inductor and said second winding of said first coupled inductor,

whereby said secondary side buck post regulator provides a second precisely regulated output with reduced magnetics volume and cost.