# Patent

## Title
Untethered device employing tunable resonant circuit

## Abstract

An untethered device is configured to inductively couple to a source device, which generates a varying magnetic field having a fixed frequency. The untethered device includes a tunable resonant circuit having a resonance frequency and configured to generate a supply voltage for the untethered device in response to the varying magnetic field. The untethered device further includes a frequency control circuit coupled to the tunable resonant circuit and configured to produce a control signal using signals developed within the tunable resonant circuit. The control signal is useable to adjust the resonance frequency of the tunable resonant circuit to the fixed frequency of the source device.

## Background
The present invention relates generally to communication between an untethered device and a sensing system and, more particularly, to sensing systems and methods that employ an untethered stylus as a user input implement.
BACKGROUND
Personal computing systems of varying type and configuration typically provide one or more user interface devices to facilitate user interaction with such computing systems. Well known user interface devices include a keyboard, mouse, trackball, joystick, and the like. Various types of personal computing devices, such as tablet PCs, provide a pen apparatus that can be manipulated by the user, much in the same way as a pencil or ink pen.
Conventional computing devices that provide for user input via a pen or other pointer implement typically employ an electromagnetic inductive system. The electromagnetic inductive system usually comprises an electromagnetic pen or pointer apparatus and a digitizer in the form of a tablet. Changes in pen location relative to the digitizer's sensing surface are detected and location computations are made to determine the coordinates of the pen.
SUMMARY OF THE INVENTION
The present invention is directed to systems and methods for enhancing communication between an untethered device and a sensing system. According to embodiments of the present invention, an untethered device is configured to inductively couple to a source device. The source device is configured to generate a varying magnetic field having a fixed frequency. The untethered device includes a tunable resonant circuit having a resonant frequency and configured to generate a supply voltage for the untethered device in response to the varying magnetic field. The untethered device further includes a frequency control circuit coupled to the tunable resonant circuit and configured to produce a control signal using signals developed within the tunable resonant circuit. The control signal is useable to adjust the resonant frequency of the tunable resonant circuit to the fixed frequency of the source device.
The tunable resonant circuit includes an inductive circuit coupled in parallel to a capacitive circuit, and the frequency control circuit is configured to produce the control signal using voltages associated with the inductive and capacitive circuits, respectively. For example, the inductive circuit may include an inductive component coupled in series to a first resistance, and the capacitive circuit may include a capacitive component coupled in parallel to a second resistance. The frequency control circuit may be configured to produce the control signal using voltages developed across the first and second resistances, respectively.
In one implementation, the tunable resonant circuit includes an inductive circuit coupled in parallel to a capacitive circuit, and the control signal produced by the frequency control circuit is useable to adjust one of the inductive circuit and the capacitive circuit. In another implementation, the tunable resonant circuit includes an inductive circuit coupled in parallel to a capacitive circuit, and the control signal produced by the frequency control circuit is useable to adjust each of the inductive and capacitive circuits.
According to another implementation, the tunable resonant circuit includes an inductive circuit coupled in parallel to a capacitive circuit, and the frequency control circuit is configured to produce the control signal using a first voltage (V1) associated with the inductive circuit and a second voltage (V2) associated with the capacitive circuit. The control signal, according to this implementation, may be defined by Vc cos φ, where Vc is proportional to the sensed voltage signal developed across the second resistance, and φ is a phase angle between the sensed voltage signals developed across the first and second resistances, respectively. The control signal, Vc cos φ may also be defined such that Vc is proportional to V1 and V2, and φ is a phase angle between V1 and V2.
The frequency control circuit may be configured to include a multiplier having a number of inputs, each of the inputs configured to receive one of the signals developed within the tunable resonant circuit. A low-pass filter may be coupled to the multiplier and configured to remove undesirable high frequency content of the signal output from the multiplier. The control signal useable to adjust the resonant frequency of the tunable resonant circuit to the fixed frequency of the source device is provided at an output of the low pass filter.
In various embodiments, the source device may include an RFID reader and the untethered device may include an RFID tag. In other embodiments, the source device includes a digitizer and the untethered device is configured as a stylus.
In accordance with other embodiments, methods of the present invention may be implemented involving an untethered device configured to inductively couple to a source device, the source device configured to generate a varying magnetic field having a fixed frequency. Methods of the present invention may involve generating, by a tunable resonant circuit of the untethered device, a supply voltage for the untethered device in response to the varying magnetic field. Signals developed within the tunable resonant circuit are sensed, the signals associated with parallel connected inductive and capacitive circuits, respectively, of the tunable resonant circuit. A control signal is produced using the sensed signals, the control signal useable to adjust a resonant frequency of the tunable resonant circuit to the fixed frequency of the source device.
Sensing signals developed within the tunable resonant circuit may involve sensing voltage signals associated with the inductive and capacitive circuits, respectively. For example, the inductive circuit may include an inductive component coupled in series to a first resistance, and the capacitive circuit may include a capacitive component coupled in parallel to a second resistance. Sensing signals developed within the tunable resonant circuit may involve sensing voltage signals developed across the first and second resistances, respectively.
In one approach, the control signal is defined by Vc cos φ, where Vc is proportional to the sensed voltage signal developed across the second resistance, and φ is a phase angle between the sensed voltage signals developed across the first and second resistances, respectively. In another approach, the control signal, Vc cos φ, may be defined such that Vc is proportional to V1 and V2, and φ is a phase angle between V1 and V2. The control signal is useable to adjust one or each of the inductive circuit and the capacitive circuit. Producing the control signal may involve performing a multiplier or comparison operation on the sensed signals, and may further involve low-pass filtering an output of the multiplier or comparison operation.
The above summary of the present invention is not intended to describe each embodiment or every implementation of the present invention. Advantages and attainments, together with a more complete understanding of the invention, will become apparent and appreciated by referring to the following detailed description and claims taken in conjunction with the accompanying drawings.



BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a diagram of a location sensing system that includes an untethered stylus and a location sensing device in accordance with embodiments of the present invention;
FIG. 2 is a diagram of various components of a location sensing device that cooperates with a stylus in accordance with embodiments of the present invention;
FIG. 3 is a diagram of an apparatus for generating an excitation magnetic field which is received by a stylus in accordance with embodiments of the present invention;
FIG. 4 is an illustration of various components of a stylus implemented in accordance with embodiments of the present invention;
FIG. 5 shows a schematic model of a parallel coil-capacitor circuit that may be incorporated in a stylus or other device, such as an RFID tag, in accordance with embodiments of the present invention;
FIG. 6 is a block diagram of one illustrative implementation that provides for continuous tuning of resonant circuitry of an untethered device, such as a stylus or RFID tag, in accordance with embodiments of the present invention;
FIG. 7 is a schematic of circuitry of a source device and an untethered device in accordance with embodiments of the present invention;
FIG. 8 is a block diagram of a frequency control circuit in accordance with embodiments of the present invention;
FIG. 9 is a plot of cos φ versus normalized resonance frequency fres/fs for different R1C time constants in connection with the circuitry shown in FIG. 7, the plot of FIG. 9 showing a desired zero value at the imposed source frequency fs in accordance with embodiments of the present invention;
FIG. 10 is a schematic of circuitry for controlling the resonance frequency of inductive-capacitive (LC) circuitry of an untethered device in accordance with embodiments of the present invention;
FIGS. 11 and 12 show the results of changing a control voltage, Vc, on the resonance frequency of LC circuitry shown in FIG. 10;
FIG. 13 is a block diagram of components that provide for self-tuning of the frequency of a drive source that inductively couples to an LC circuit of an untethered device, such as a stylus or RFID tag, in accordance with embodiments of the present invention;
FIG. 14 is a schematic of circuitry that may be incorporated in the source device and untethered device shown in FIG. 13;
FIG. 15 is a plot of capacitor voltage and input impedance phase as a function of frequency in connection with the circuitry depicted in FIGS. 13 and 14; and
FIG. 16 is a plot of the phase of source device input impedance, Zin, for different values of k in connection with the circuitry depicted in FIGS. 13 and 14.



While the invention is amenable to various modifications and alternative forms, specifics thereof have been shown by way of example in the drawings and will be described in detail. It is to be understood, however, that the intention is not to limit the invention to the particular embodiments described. On the contrary, the intention is to cover all modifications, equivalents, and alternatives falling within the scope of the invention as defined by the appended claims.
DETAILED DESCRIPTION OF EMBODIMENTS
In the following description of the illustrated embodiments, reference is made to the accompanying drawings which form a part hereof, and in which is shown by way of illustration, various embodiments in which the invention may be practiced. It is to be understood that the embodiments may be utilized and structural changes may be made without departing from the scope of the present invention.
The present invention is directed to methods and systems for effecting communication between an untethered device and a sensing system. Embodiments of the present invention provide for enhanced communication of information, such as analog and/or digital information, between the untethered device and the sensing system. Exemplary embodiments of the untethered device include a stylus or RFID tag. Exemplary embodiments of the sensing system include a device locating sensor or touch location sensor, such as a digitizer, or a RFID reader. Other examples of untethered devices include game pieces that interacts with a game board or other structure that incorporates a sensing system. It is understood that methods and systems in accordance with the present invention may be implemented in a wide range devices and systems other than those particularly described herein.
Embodiments of the present invention are directed to tuning of resonant circuitry provided at an untethered device, such as a stylus or RFID tag. Implementations of such embodiments provide for optimal continuous tuning of resonant circuitry in an untethered device, such as by extracting an error control signal that can be used in a feedback mechanism to self-tune the resonance frequency of inductive-capacitive circuits. In this manner, the need for time consuming hand tuning of individual circuits at the manufacturing level is eliminated, and optimal performance under varying external conditions, such as temperature, is achieved.
Other embodiments of the present invention are directed to controlling the resonance frequency of LC circuits in an untethered device using a control voltage to change the capacitance, C, of this LC circuit. An automatic mechanism using feedback according to such embodiments provides for precise and continuous tuning of the capacitor value, C, in order to ensure optimal circuit performance. Such a mechanism obviates time consuming and expensive manual tuning of component values and subsequent detuning because of slowly changing conditions.
According to further embodiments of the present invention, methodologies provide for self-tuning the frequency of a source device, so as to optimize an induced voltage over a capacitor in an LC circuit of an untethered device. Such embodiments provide methodologies for self-tuning the frequency of a driver that inductively couples to the LC circuit of an untethered device so that a maximum voltage is developed over the capacitor C in the LC circuit.
Embodiments of an untethered stylus of the present invention may be implemented in the context of a location sensing system, embodiments of which are illustrated in FIGS. 1-3. According to the embodiments shown in FIGS. 1-3, a location sensing system 10 includes a stylus 12 that interacts with a sensing device 11. The sensing device 11 includes a location sensor 14, such as a digitizer. The stylus 12 is preferably configured as a tetherless or cordless implement that does not have a battery. Rather, the stylus 12 derives power from a magnetic field generated by the sensing device 11. Although preferred embodiments of an untethered stylus do not include a battery, some embodiments may employ a battery, such as a rechargeable battery that is recharged from energy derived from the magnetic field of the drive signal. A battery may be used to provide power to various circuits of the stylus, such as a modulator or pressure sensor (e.g., tip or eraser pressure sensor).
The sensing device 11 is shown to include a drive loop or coil 18 coupled to drive loop electronics 16 that cooperate to generate a magnetic field, which may be a continuously varying magnetic field. Drive coil 18 may comprise one or more coils or loops. The stylus 12, having derived power from the magnetic field emanating from the drive coil 18, broadcasts a signal from which stylus location and status may be determined by the sensing device 11.
The stylus 12 is preferably configured to include one or more user-actuatable buttons or switches, such as those commonly employed to implement various mouse functions (e.g., right and left mouse buttons). The tip of the stylus 12 may incorporate a pressure sensor from which applied pressure can be resolved and transmitted to the sensing device 11. Eraser functionality may also be incorporated in the form of a switch or pressure sensor at the stylus end opposite the tip.
Sensor interface electronics 20 is coupled to the sensor 14 and facilitates measurement of signals developed at the sensor 14 in response to signals broadcast by the stylus 12. According to one configuration, the sensor 14 includes a digitizer that incorporates a detection grid and electronics as is known in the art. For example, such a detection grid may include pairs of position resolving conductors each of which forms one or more differential coil elements in the sensor 14, with each conductor pair receiving a magnetic signal transmitted by the stylus 14. An illustrative example of a digitizer having such a detection grid configuration, elements of which may be employed in a location sensor system of the present invention, is disclosed in U.S. Pat. Nos. 4,786,765; 5,218,174; 5,633,471; 5,793,360; 6,667,740; and 7,019,672; which are hereby incorporated herein by reference.
According to another configuration, the sensing device 11 may incorporate a sensor 14 that effectively incorporates a digitizer and a touch-sensitive sensor. The digitizer, according to this configuration, allows the location and status of the stylus 12 to be determined. The touch-sensitive sensor allows the location of a finger touch to be determined. This configuration allows a user to use either the stylus 12 or a finger to indicate a desired location on a computer display, as well as determine the location and status of the stylus 12.
The touch-sensitive sensor 14 typically includes a matrix that capacitively couples to the stylus 12 and/or a finger. In this configuration, the sensor 14 of the sensing device 11 is preferably made up of a series of transparent conductors placed upon a glass or plastic cover that can be placed in front of an LCD display. One side of the glass or plastic sheet has conductors in the X direction, and the opposite side has conductors in the Y direction. Examples of suitable touch sensitive sensors 14 are disclosed in commonly owned U.S. Pat. Nos. 6,133,906 and 6,970,160, in commonly owned U.S. Published Application No. 2005/0083307, in U.S. Pat. Nos. 6,762,752 and 6,690,156, and in U.S. Published Application No. 2004/0095333, each of which is hereby incorporated herein by reference.
An embodiment that incorporates a digitizer and touch-sensitive sensor advantageously allows a user to point a stylus at a computer display and have the location and status of the pointing device determined and, when a finger is used to point at the display device, allows for the determination of the location of a finger touch at the display device. The dual use aspects of this embodiment of a sensing device 11 make it particularly useful in tablet PC applications.
For example, a digitizer arrangement allows a user to use a stylus to input information, indicate operations the user wants to take, and write or draw on the display. The touch-sensitive sensor allows the user to “type” information onto a virtual keyboard on the display screen, for example. This would allow the vendor of the computing system, in which a dual touch location sensor system of the present invention is implemented, to eliminate the keyboard and the associated bulk it requires. It is understood that a digitizer and a touch-sensitive sensor need not be implemented together in all configurations, but inclusion of both sensing devices provides for enhanced user interaction with a computing system that incorporates a sensing system 10 of the present invention.
According to one embodiment, the drive coil 18 may be constructed of wire, such as 36 gauge wire, looped several times (e.g., 4 times) around the periphery of the frame of sensing device 11. In one implementation, the drive coil 18 may have an inductance of about 21 μH and an impedance of about 14 Ohms at 100 kHz. The drive coil 18 is connected to a signal generator of the drive loop electronics 16. The signal generator may be configured to produce 200 periods of a 100 kHz sine wave signal gated at 250 Hz. The signal generator may, for example, produce an output signal of 0.4 Vpp, resulting in approximately 28 mA of current that flows in the drive coil 18.
FIG. 3 is a simplified illustration of drive coil 18 and a signal generator 17 that cooperate to generate a magnetic excitation field. In this illustrative example, one or more coils are preferably arranged in the plane of the location sensor. A sinusoidal current is produced by the signal generator 17 with peak magnitude A1 at radian frequency ω1 and is applied to the rectangular coil 18.
The stylus 12 is configured to collect energy from the magnetic field generated by drive coil 18/drive loop electronics 16 using a tank circuit. The tank circuit is preferably tuned to resonate at the frequency that the drive coil 18 is driven. In this illustrative example, the frequency is set at 100 kHz. The tank circuit of the stylus 12 builds amplitude during the burst produced by the drive coil 18 and then gradually loses signal amplitude after the drive coil 18 is turned off. The time associated with the exponential charging and discharging of the resonant tank circuit of the stylus 12 is determined by the capacitive and inductive elements in the tank circuit. Matching of the tank circuit's resonance frequency and the drive signal frequency is preferably accomplished using one or more of the techniques described hereinbelow.
Referring again to FIG. 1, the sensor interface electronics 20 is preferably connected to the sensor 14 via a shielded connector. The sensor interface electronics 20 includes circuitry for measuring the signal levels present on the individual traces of the sensor 14, and is typically configured to reject as much noise as possible.
As is shown in FIG. 2, an envelope detector circuit 30 of the sensor interface electronics 20 is configured to detect signals developed on individual traces of the sensor 14. The signals output by the envelope detector circuit 30 are digitized by use of analog-to-digital (A/D) converters 32. Each trace of the sensor 14 may have a dedicated A/D converter 32. Alternatively, two or more traces may share a common A/D converter 32 via a switch having a sufficient switching frequency. The envelope detector circuit 30 is configured to provide sufficient gain to make the resultant signal match the requirements of A/D converters 32. The envelope detector circuit 30 may be configured to generate a signal having the same shape as an imaginary line describing the upper bound of the sensor signal. In such a configuration, the envelope detector circuit 30 effectively transforms the 100 kHz signal into a DC or low frequency signal that is more readily digitized. The envelope detector circuit 30 preferably incorporates one or more synchronous demodulators.
A processor 22 is coupled to the drive loop electronics 16, sensor interface electronics 20, and a communications interface 24, as is shown in FIG. 1. The processor 22 coordinates the operations of drive loop electronics 16 and sensor interface electronics 20, and is configured to determine stylus/finger location and stylus status. Stylus/finger location and stylus status determinations may be made by the processor 22 using known approaches, such as those discussed in the patent references incorporated herein by reference. In one embodiment, processor 22 determines stylus/finger location and stylus status in accordance with the methodologies disclosed in commonly owned U.S. patent application Ser. No. 11/557,829, entitled “Touch Location Sensing System and Method Employing Sensor Data Fitting to a Predefined Curve,” filed on Nov. 8, 2006, which is hereby incorporated herein by reference.
The location and status information computed by the processor 22 is communicated to a computer and/or display 26 via a communications interface 24. The communications interface 24 may be configured as an RS-232 or USB interface, for example. The processor 22 may be configured to drive a display 26 directly. Alternatively, a computer 28 may be coupled to the communications interface 24 and receive the location and status information from the processor 22, and drive its display. The processor 22 or computer 28 may be configured to control cursor velocity, momentum and other factors to enhance the user experience with the sensing system 11.
Referring now to FIG. 4, there is shown an embodiment of an untethered stylus 12 of the present invention that may be implemented in the context of a location sensing system as described above or other sensing system known in the art. In accordance with the embodiment shown in FIG. 4, a stylus 12 houses electronics 52, which includes frequency sensitive LC circuitry, and a coil 54 wrapped around a ferrite cylinder 53. The ferrite cylinder 53 serves to increase signal amplitude. An applied harmonic magnetic field produced at the surface of the location sensor (e.g., digitizer) or a display, for example, couples flux through the ferrite cylinder 53 and thus to the coil 54 when the stylus 12 is placed in the applied field.
The ferrite coil arrangement 56 resonates with a separate parallel-connected capacitor of the electronics 52 and is tuned to the excitation field frequency. In various embodiments, circuitry 55 is provided in the untethered stylus 12 to provide for self-tuning of the resonance frequency of the frequency sensitive LC circuitry in accordance with one or more techniques described herein. In other embodiments, the frequency of a source voltage produced by the source device that generates the excitation field is adjusted so that the phase of an input impedance of the source device is substantially zero, thereby maximizing a voltage across the capacitor of the frequency sensitive LC circuitry of the untethered stylus 12.
The parallel coil-capacitor combination is connected between the stylus tip 57 and the stylus shield 59. The shield 59 may form part of, or otherwise be connected to, the stylus housing so that it can be touched, and therefore grounded, by a user's hand when held. The shield 59 may be situated to extend over the circuitry region of the stylus 12, and preferably has a discontinuous shape, such as a “C” shape, so as to avoid eddy currents that could otherwise arise in a closed loop shield arrangement.
The stylus tip 57 couples capacitively to the location sensor from which location information is derived. To provide stylus status information, the ferrite coil arrangement 56 powers the electronics 52 which amplitude modulates the stylus tip voltage at the reference frequency or frequencies. The frequency of the oscillations is changed to reflect the stylus status, such as switch closures or tip pressure changes.
Alternatively, the invention may be implemented with magnetic-sensing digitizer systems as are known in the art. An untethered magnetic stylus is similar to the capacitive stylus shown in FIG. 4, except the resonant circuit comprising ferrite coil arrangement 56 and separate parallel-connected capacitor of the electronics 52 need not be connected to tip 57 nor to a shield 59. Untethered magnetic styluses are well known in the art, and are described in previously incorporated U.S. Pat. Nos. 4,786,765; 5,633,471; 5,793,360; 6,667,740, and 7,019,672. Embodiments of the present invention that are implemented using an untethered magnetic stylus may employ a location sensor that includes multiple drive loops as disclosed in the referenced patents. In such embodiments, a separate sensing grid and separate drive loops need not used. Rather, each of the drive loop coils is alternately coupled to transmitting circuitry and then to receiving circuitry to alternately transmit and receive from one of multiple drive loop coils that are placed in the active area, typically under the display.
FIG. 5 shows a schematic model of a parallel coil-capacitor circuit that facilitates an enhanced understanding of the present invention. The parallel coil-capacitor circuit shown in FIG. 5 (i.e., tank circuit) may be incorporated in a stylus as part of, or coupled to, resonance frequency tuning circuitry in accordance with embodiments of the present invention. FIG. 5 shows a capacitor C1 connected in parallel with a coil 54 to resonate at the excitation frequency or the transmitted frequency. The voltage developed across the coil 54, which is shown modeled as voltage generator 61, is coupled to the stylus tip 57 and then capacitively coupled to the location sensor, such as sensor 14 shown in FIG. 1. The voltage developed across the resonating coil 54 is preferably modulated with one or a combination of known techniques. An added ferrite cylinder 53 about which coil 54 is preferably wrapped, as shown in FIG. 5, has the effect of increasing the magnetic flux B and signal coupled by the drive coil of the location sensor to the receiving coil 54 of the stylus 12.
The capacitance value of capacitor C1 shown in FIG. 5 is selected such that the capacitance, C, of capacitor C1 resonates with the coil inductance, L, at the excitation angular frequency ω so that there is no voltage drop across the LC combination. Two different voltages in this circuit can be considered. The first voltage of consideration is the voltage V (shown in terms of voltage source 61) that develops across the coil 54 through magnetic induction. It is well understood that this voltage 61 is basically equal to the number of stylus coil turns N times the coil cross section A times the rate of change of the magnetic flux density passing through the ferrite cylinder, which is given by V=N*A*dB/dt.
The second voltage of consideration is the voltage that develops across the capacitor C1. This voltage VC is also the stylus tip voltage. From basic circuit analysis at resonance, it follows that: VC=V/(ωRC)=V(ωL/R) with the quantity 1/(ωRC)=(Lω)/R defined as the resonant circuit quality factor Q, where ω is expressed in terms of radians per second. This second voltage may be modulated for purposes of communicating stylus status data to a location sensor.
With continued reference to FIG. 5, one approach to transmitting stylus status information in addition to stylus position information is through addition of a second capacitor C2 connected to the first capacitor C1 through a switch 16. Opening and closing the switch 16 causes the resonance frequency of the coil-capacitor combination 54/C1 to change. This change may be detected by observing a change in phase of the stylus transmitted frequency or though a transient frequency change caused when the drive coil current is turned off.
This method of data transmission, however, is not suitable for a stylus powered by a constantly varying magnetic field and capacitively coupled to the digitizer. Constant excitation does not allow a transient measurement of the stylus resonance, and phase modulation is difficult to detect as the phase of the digitizer received signal varies dramatically as the stylus is moved across the location sensor (e.g., digitizer). Frequency modulation of an amplitude-modulated signal using multiple reference frequencies generated at the stylus, for example, removes these difficulties. The location sensor may be configured to demodulate the amplitude modulation and detect the reference frequencies of the modulation.
FIG. 6 is a block diagram of one illustrative implementation that provides for continuous tuning of resonant circuitry of an untethered device 12, such as a stylus or RFID tag. According to this illustrative embodiment, the untethered device 12 implements a method of extracting an error control signal that can be used in a feedback mechanism to self-tune the resonance frequency of LC circuitry of the untethered device 12.
According to the implementation shown in FIG. 6, an untethered device 12 includes a tunable resonant circuit 62, which includes inductive and capacitive circuit components. The tunable resonant circuit 62 has a resonance frequency that can be continuously adjusted to match the frequency of the source device 60. The source device 60 typically generates a varying magnetic field having a fixed frequency.
The untethered device 12 is further shown to include a frequency control circuit 64, which is coupled to the tunable resonant circuit 62 via a control line 66. The frequency control circuit 64 is configured to produce a control signal, Sc, using signals developed within the tunable resonant circuit 62. In this illustrative example, voltage signals V1 and V2, developed within the tunable resonant circuit 62, are extracted for use in tuning the tunable resonant circuit 62. The control signal, SC, generated by the frequency control circuit 64, is useable to adjust the resonance frequency of the tunable resonant circuit 62 to match the fixed frequency of the source device 60.
Turning now to FIG. 7, there is shown a schematic of circuitry of a source device 60 and an untethered device 12 in accordance with embodiments of the present invention. In configurations in which the untethered device 12 is implemented as an RFID tag, the L1-C circuit 72, 74 is referred to as the “tag,” while the L0 inductance component 71 is part of a “reader,” with fixed frequency fs. It is understood that the circuitry shown in FIG. 7 and in other Figures may be employed in systems and devices other than those that employ untethered styli and RFID tags.
In conventional implementations, the components L1 72 and C 74 are fixed, such that the resonance frequency, fres, of the L1-C combination 72, 74 is defined as:




  
    f
    res
  
  =
  
    1
    
      2
      ⁢
      π
      ⁢
      
        
          
            L
            1
          
          ⁢
          C
        
      
    
  




The resonance frequency, fres, of the L1-C combination 72, 74 is not exactly equal to the source frequency, fs. As a result, the voltage V2 that develops across the capacitor C 74 is lower than desired. Because this voltage, V2, is often used to power additional circuitry or communication electronics, a reduced voltage across the capacitor C 74 results in suboptimal performance. A tunable resonant circuit with frequency control in accordance with the present invention provides an improvement in performance, in that L1 72, C 74 or both may be automatically and continuously tuned so that the resonance frequency, fres, of the L1-C combination 72, 74 becomes exactly equal to the frequency, fs, of the source device 60.
Defining the angular resonance frequency ωres=2πfres and applying Kirchoff's laws to the circuit in FIG. 7 leads to the following:




  
    
      
        V
        2
      
      
        V
        1
      
    
    ⁢
    
      
        (
        
          
            ω
            2
          
          -
          
            ω
            res
            2
          
        
        )
      
      
        ω
        res
        2
      
    
    ⁢
    
      
        R
        2
      
      
        R
        1
      
    
  
  -
  
    jω
    ⁢
    
      
    
    ⁢
    
      R
      2
    
    ⁢
    C
  




so that the phase angle, φ between V1 and V2 is given by:




  φ
  =
  
    
      arctan
      ⁢
      
        
      
      ⁢
      
        
          ω
          res
          2
        
        
          
            ω
            res
            2
          
          -
          
            ω
            2
          
        
      
      ⁢
      ω
      ⁢
      
        
      
      ⁢
      
        R
        1
      
      ⁢
      C
    
    =
    
      arctan
      ⁢
      
        
      
      ⁢
      
        
          2
          ⁢
          π
          ⁢
          
            
          
          ⁢
          
            fR
            1
          
          ⁢
          C
        
        
          (
          
            1
            -
            
              
                (
                
                  f
                  /
                  
                    f
                    res
                  
                
                )
              
              2
            
          
          )
        
      
    
  




Voltage signals V1 and V2 developed as shown in FIG. 7 are preferably fed to a frequency control circuit, a general embodiment of which is shown in FIG. 6. Using the voltage signals V1 and V2 as inputs, the frequency control circuit produces a control signal that is fed back to the resonant circuitry which, in response, results in adjustment of the resonance frequency of the untethered device 12 to match the frequency of the source voltage produced by the source device 60.
In general, the control signal generated by the frequency control circuit is preferably a signal that is proportional to the voltage signals V1 and V2. The control signal may also be a signal that is proportional to the voltage signal V2. According to one approach, the control signal is preferably produced by a multiplication of the voltage signals V1 and V2. If V1 and V2 are harmonic and have the same frequency but a phase shift of φ, as defined above, the multiplication results in the following:




  
    
      
        
          V
          out
        
        =
        
          
            V
            1
          
          ⁢
          cos
          ⁢
          
            
          
          ⁢
          ω
          ⁢
          
            
          
          ⁢
          t
          *
          
            V
            2
          
          ⁢
          
            cos
            ⁡
            
              (
              
                
                  ω
                  ⁢
                  
                    
                  
                  ⁢
                  t
                
                -
                φ
              
              )
            
          
        
      
    
  
  
    
      
        =
        
          
            
              
                V
                1
              
              ⁢
              
                V
                2
              
            
            2
          
          ⁢
          
            {
            
              
                cos
                ⁢
                
                  
                
                ⁢
                φ
              
              +
              
                cos
                ⁡
                
                  (
                  
                    
                      2
                      ⁢
                      ω
                      ⁢
                      
                        
                      
                      ⁢
                      t
                    
                    -
                    φ
                  
                  )
                
              
            
            }
          
        
      
    
  




The first term in the brackets of the resultant equation, cos φ, is the desired control signal component. The second term, cos(2ωt−φ), is a double frequency signal component that is undesirable. This second term is preferably filtered out of the control signal, which, after filtering, can be represented as Vc cos φ. Those skilled in the art will appreciate that various known techniques used in a variety of mixing and superheterodyning applications may be employed to extract a control signal usable to adjust the resonance frequency of the tunable resonant circuit of the untethered device 12 to match the fixed frequency of the source device 60 in accordance with the present invention.
FIG. 8 is a diagram of one implementation of a frequency control circuit in accordance with embodiments of the present invention. The frequency control circuit shown in FIG. 8 includes a multiplier 93 coupled to a low pass filter 95. The voltage signals V1 and V2, such as those shown in FIGS. 6 and 7, are fed into the multiplier 93. The output of the multiplier 93, Vout, is a voltage signal produced by a direct multiplication of V1 and V2, such that Vout=V1*V2.
As discussed above, if V1 and V2 are harmonic and have the same frequency but a phase shift of φ, then Vout is calculated according to the equation above. This signal, Vout, is then fed to the low pass filter 95, which may be configured to filter out the double frequency signal component discussed above or to simply pass only a DC signal. The output of low pass filter 95 is the control signal Vc cos φ. It is noted that both V1 and V2 will not vanish near the resonance frequency, fres, of the untethered device's tunable resonant circuit, so that Vc will always have an acceptable amplitude.
In low frequency applications (e.g., <1 MHz), the multiplier 93 may include an analog multiplier chip, such as a four-quadrant analog multiplier model AD633 available from Analog Devices, Inc. For high frequency applications (i.e., >1 MHz), any suitable mixer may be used.
FIG. 9 is a plot of cos φ versus normalized resonance frequency fres/fs for different R1C time constants in connection with the frequency control circuit shown in FIG. 8. The plot of FIG. 9 clearly shows a desired zero value at the imposed source frequency fs. Thus, the signal Vc cos φ can be used as a control signal to force the L1-C circuit shown in FIG. 7 to adjust fres to become equal to fs through a feedback mechanism. FIG. 9 shows that the dynamic range of cos φ is greater for smaller values of R1 and C. A small value of R1 is consistent with the requirement that the resonance circuit has a high Q value.
FIG. 10 is a schematic of circuitry for controlling the resonance frequency of LC circuitry of an untethered device in accordance with other embodiments of the present invention. Control circuitry implemented in accordance with embodiments encompassed by FIG. 10 provides for controlling the resonance frequency of LC circuitry of an untethered device using a control voltage that changes the capacitance, C, of the LC circuitry.
It is understood by those skilled in the art that inductances (L) and capacitances (C), as well as other factors, influence the resonance frequency of a circuit made up of these components. For high quality circuits, for example, resonance frequency variation can be very sensitive for even slightly different component values. To avoid time consuming and thus expensive manual tuning of component values, and to avoid subsequent detuning because of slowly changing conditions, an automatic mechanism using feedback in accordance with embodiments of the present invention provides for precise continuous tuning of the capacitor value, C, to ensure optimal circuit performance.
FIG. 10 shows one configuration of a voltage controlled resonance frequency LC circuit in accordance with embodiments of the present invention. In the implementation shown in FIG. 10, a coil 79 represents an inductance L of the circuit, and C1 81 and C2 82 are varactor diodes that are always driven in reverse so that they behave as capacitances. An output of an OPAMP 80 is shown coupled to the center tap of the coil 79. In this arrangement, the OPAMP output voltage as the center tap voltage for coil 79 placed both varactor diodes 81, 82 in reverse bias with the same voltage, so that both Vb and Vc can control the resonance frequency of the L, C1, C2 combination.
The bias voltage, Vb, is essentially DC and the control voltage, Vc is at most slowly varying compared to the resonance frequency of the LC circuit. The OPAMP output voltage, Vo, is given by:




  
    V
    o
  
  =
  
    
      
        -
        
          
            R
            3
          
          
            R
            1
          
        
      
      ⁢
      
        V
        c
      
    
    -
    
      
        
          R
          3
        
        
          R
          2
        
      
      ⁢
      
        V
        b
      
    
  




Demanding that varactor diodes C1 81 and C2 82 are always driven in reverse or that the OPAMP output voltage, Vo, is always negative, as given below:

−Vs<Vo<0

results in the following allowed operating range for the control voltage, Vc:





  
    
      -
      
        
          R
          1
        
        
          R
          2
        
      
    
    ⁢
    
      V
      b
    
  
  <
  
    V
    c
  
  <
  
    
      
        
          R
          1
        
        
          R
          3
        
      
      ⁢
      
        V
        s
      
    
    -
    
      
        
          R
          1
        
        
          R
          2
        
      
      ⁢
      
        V
        b
      
    
  




When R1 70=R2 77=R3 78, and with Vb=0.5Vs, the control voltage, Vc, has the following desirable range:





  
    -
    
      
        V
        s
      
      2
    
  
  <
  
    V
    c
  
  <
  
    
      V
      s
    
    2
  




Because of the anti-series arrangement of the varactor diodes C1 81 and C2 82, an external alternating flux that is coupled by coil 79 will induce a voltage over coil 79 that will cause varactor diode C1 81 to be driven less in reverse and varactor diode C2 82 more in reverse. Assuming a linear varactor capacitance-voltage dependence ΔC=kΔV, this means that varactor diode C1 81 undergoes a capacitance change +ΔC whereas varactor diode C2 82 undergoes a −ΔC change. With C1=C2=C, the total capacitance, Ct, formed by C1 81 and C2 82 is then given by:




  
    1
    
      C
      t
    
  
  =
  
    
      1
      
        C
        +
        
          Δ
          ⁢
          
            
          
          ⁢
          C
        
      
    
    +
    
      1
      
        C
        -
        
          Δ
          ⁢
          
            
          
          ⁢
          C
        
      
    
  




so that:





  
    C
    t
  
  =
  
    
      C
      2
    
    -
    
      
        
          (
          
            k
            ⁢
            
              
            
            ⁢
            Δ
            ⁢
            
              
            
            ⁢
            V
          
          )
        
        2
      
      
        2
        ⁢
        C
      
    
  




meaning that the total capacitance, Ct, is virtually AC voltage amplitude independent, except for a second order effect.

In one implementation, a total of 8 varactor diodes as 2 groups of 4 parallel diodes may be used to form C1 81 and C2 82. A ferrite containing self-inductance (L) coil 79 of 22 mH may be used so that the total capacitance, Ct=C1C2/(C1+C2), at Vb=7.5 V and Vc=0 V is estimated to be about 115 pF. Therefore, C1 81 and C2 82 each have a capacitance of approximately 230 pF. The sensitivity for such an implementation is around 1 kHz/V.
The circuit in FIG. 10 may be implemented using silicon varactor diodes, such as MVAM115 silicon varactor diodes available from Advanced Semiconductor, Inc. OPAMP 80 is preferably a low noise OPAMP, such as OP27 Low Noise, Precision Operational Amplifier available from Analog Devices, Inc.
FIGS. 11 and 12 show the results of changing control voltage, Vc, on the resonance frequency. FIG. 11 shows a resonance peak for Vc=0 volt, at a resonance frequency≈100 kHz. FIG. 12 shows a resonance peak for Vc=5 volt, at a resonance frequency≈95 kHz. The frequency spectra in FIGS. 11 and 12 were obtained by inductively inducing a white spectrum into coil 79 and measuring the LC circuit response after changing Vc.
Further embodiments of the present invention are directed to self-tuning the frequency of a drive source that inductively couples to an LC circuit of an untethered device, such as a stylus or RFID tag, so that a maximum voltage develops over the capacitor, C, in the LC circuit. FIG. 13 illustrates an implementation of one such embodiment, in which a source device 60 inductively couples to an LC circuit in an untethered device 12. The source device 60 is shown to include a driver circuit 92 coupled to drive coil 18, which may include one or more coils. The driver circuit 92 may be implemented as part of the drive loop electronics 16 shown in FIG. 1.
Also shown in FIG. 13 is a phase detector 96 coupled to the driver circuit 92, the driver circuit 92 further including a drive coil circuit. A controller 94 is coupled to the phase detector 96 and the driver circuit 92. The controller 94 may be part of, or coupled to, the processor 22 shown in FIGS. 1 and 2.
In operation, the driver circuit 92 and drive coil 18 cooperate to generate a varying magnetic field. The phase detector 96 detects a phase of an input impedance of the driver circuit 92 in response to the source device 60 inductively coupling with the LC circuit of the untethered device 12. The controller 94 adjusts a frequency of a source voltage applied to the driver circuit 92 in response to an output signal of the phase detector 96. The controller 94, which may be implemented using a voltage controlled oscillator, adjusts the source voltage frequency so that the phase of the input impedance as indicated by the output signal of the phase detector 96 is substantially zero. In this manner, controller adjustment of the source voltage frequency compensates for a change in the resonance frequency, or a parameter of the resonance frequency, of the resonant LC circuit of the untethered device 12.
FIG. 14 is a schematic of circuitry that may be incorporated in the source device 60 and untethered device 12 shown in FIG. 13. FIG. 14 shows the source device 60 inductively coupled to an LC resonance circuit of the untethered device 12. The frequency matching methodology according to this embodiment involves detecting the phase of the source device's input impedance, Zin (looking into the drive coil 84 (L1)), such as by use of a phase detector, and adjusting the frequency of the source voltage V 65 until that phase is zero. The output of the phase detector can then be used in a feedback circuit to drive a voltage controlled oscillator (VCO), for example, with V as an output signal.
A variety of available phase detectors may be used to perform phase detection and comparison. In low frequency applications (i.e. <1 MHz), for example, an analog multiplier, such as the aforementioned AD633, may be used. In higher frequency applications (i.e., >1 MHz), a passive component, such as a PDP-201 Phase Detector available from Synergy Microwave Corporation, may be used for phase detection. Other commercial components and implementation may be used that provide similar functionality and performance.
The following analysis is provided with reference to FIG. 14. With sufficient coupling k, the phase of Zin is zero at the frequency that corresponds with the maximum voltage Vc. Applying Kirchoff's laws to the circuit in FIG. 14 leads to:




  
    
      
        
          
            V
            c
          
          V
        
        =
        
          
            1
            C
          
          ·
          
            
              k
              ⁢
              
                
                  
                    L
                    1
                  
                  ⁢
                  
                    L
                    2
                  
                
              
            
            
              
                
                  (
                  
                    
                      R
                      1
                    
                    +
                    
                      jω
                      ⁢
                      
                        
                      
                      ⁢
                      
                        L
                        1
                      
                    
                  
                  )
                
                ⁢
                
                  (
                  
                    
                      R
                      2
                    
                    +
                    
                      jω
                      ⁢
                      
                        
                      
                      ⁢
                      
                        L
                        2
                      
                    
                    -
                    
                      j
                      
                        ω
                        ⁢
                        
                          
                        
                        ⁢
                        C
                      
                    
                  
                  )
                
              
              +
              
                
                  ω
                  2
                
                ⁢
                
                  k
                  2
                
                ⁢
                
                  L
                  1
                
                ⁢
                
                  L
                  2
                
              
            
          
        
      
    
    
      
        [
        1
        ]
      
    
  




with ω angular frequency, whereas Zin is given by:





  
    
      
        
          Z
          
            i
            ⁢
            
              
            
            ⁢
            n
          
        
        =
        
          
            R
            1
          
          +
          
            jω
            ⁢
            
              
            
            ⁢
            
              L
              1
            
          
          +
          
            
              
                k
                2
              
              ⁢
              
                ω
                2
              
              ⁢
              
                L
                1
              
              ⁢
              
                L
                2
              
            
            
              
                R
                2
              
              +
              
                jω
                ⁢
                
                  
                
                ⁢
                
                  L
                  2
                
              
              -
              
                j
                
                  ω
                  ⁢
                  
                    
                  
                  ⁢
                  C
                
              
            
          
        
      
    
    
      
        [
        2
        ]
      
    
  




FIG. 15 is a plot of capacitor voltage and input impedance phase as a function of frequency. More particularly, FIG. 15 is a plot of the voltage ratio Vc/V of equation [1] and phase, Zin, of equation [2] for the following values of the components shown in FIG. 14: L1=45 μH, L2=22 mH, R1=9Ω, R2=40Ω, C 87=153.5 pF, and k=0.5. As is clear in FIG. 15, the capacitor voltage, Vc, is maximum near the higher frequency point where the phase of the input impedance, Zin, is zero.
The source frequency self-tuning methodology described above works well for strong coupling or high values of k, where k takes on values between 0 and 1 (i.e., 0<k<1). FIG. 16 gives an indication of the phase of Zin for different values of k. It can be seen from FIG. 16 that the detection mechanism breaks down when k gets too small because the phase of Zin no longer goes through zero. The fact that the phase of Zin for k=0.1 fails to go through zero in FIG. 16 may not be entirely obvious, but this fact can easily be confirmed by reproducing the graph depicted in FIG. 16 using Equation [2] above.
The foregoing description of the various embodiments of the invention has been presented for the purposes of illustration and description. It is not intended to be exhaustive or to limit the invention to the precise form disclosed. Many modifications and variations are possible in light of the above teaching. It is intended that the scope of the invention be limited not by this detailed description, but rather by the claims appended hereto.

## Claims

1. An untethered device configured to inductively couple to a source device, the source device configured to generate a varying magnetic field having a fixed frequency, the untethered device comprising:
a tunable resonant circuit having a resonance frequency and configured to generate a supply voltage for the untethered device in response to the varying magnetic field; and
a frequency control circuit coupled to the tunable resonant circuit and configured to produce a control signal using signals developed within the tunable resonant circuit, the control signal useable to adjust the resonance frequency of the tunable resonant circuit to the fixed frequency of the source device.

2. The device of claim 1, wherein:
the tunable resonant circuit comprises an inductive circuit coupled in parallel to a capacitive circuit; and
the frequency control circuit is configured to produce the control signal using voltages associated with the inductive and capacitive circuits, respectively.

3. The device of claim 1, wherein:
the tunable resonant circuit comprises an inductive circuit coupled in parallel to a capacitive circuit, the inductive circuit comprising an inductive component coupled in series to a first resistance, and the capacitive circuit comprising a capacitive component coupled in parallel to a second resistance; and
the frequency control circuit is configured to produce the control signal using voltages developed across the first and second resistances, respectively.

4. The device of claim 1, wherein the tunable resonant circuit comprises an inductive circuit coupled in parallel to a capacitive circuit, and the control signal produced by the frequency control circuit is useable to adjust one of the inductive circuit and the capacitive circuit.

5. The device of claim 1, wherein the tunable resonant circuit comprises an inductive circuit coupled in parallel to a capacitive circuit, and the control signal produced by the frequency control circuit is useable to adjust each of the inductive circuit and the capacitive circuit.

6. The device of claim 1, wherein:
the tunable resonant circuit comprises an inductive circuit coupled in parallel to a capacitive circuit;
the frequency control circuit is configured to produce the control signal using a first voltage (V1) associated with the inductive circuit and a second voltage (V2) associated with the capacitive circuit; and
the control signal is defined by Vc cos φ, where Vc is proportional to V1 and V2, and φ is a phase angle between V1 and V2.

7. The device of claim 1, wherein the frequency control circuit comprises a multiplier having a plurality of inputs, each of the plurality of inputs configured to receive one of the signals developed within the tunable resonant circuit.

8. The device of claim 7, comprising a low-pass filter coupled to the multiplier, the control signal provided at an output of the low pass filter.

9. The device of claim 1, wherein the source device comprises an RFID reader and the untethered device comprises an RFID tag.

10. The device of claim 1, wherein the source device comprises a digitizer and the untethered device comprises a stylus.

11. A method implemented in an untethered device configured to inductively couple to a source device, the source device configured to generate a varying magnetic field having a fixed frequency, the method comprising:
generating, by a tunable resonant circuit of the untethered device, a supply voltage for the untethered device in response to the varying magnetic field;
sensing signals developed within the tunable resonant circuit, the signals associated with parallel connected inductive and capacitive circuits, respectively, of the tunable resonant circuit; and
producing a control signal using the sensed signals, the control signal useable to adjust a resonance frequency of the tunable resonant circuit to the fixed frequency of the source device.

12. The method of claim 11, wherein sensing signals developed within the tunable resonant circuit comprises sensing voltage signals associated with the inductive and capacitive circuits, respectively.

13. The method of claim 12, wherein:
the inductive circuit comprises an inductive component coupled in series to a first resistance, and the capacitive circuit comprises a capacitive component coupled in parallel to a second resistance; and
sensing signals developed within the tunable resonant circuit comprises sensing voltage signals developed across the first and second resistances, respectively.

14. The method of claim 13, wherein the control signal is defined by Vc cosφ, where Vc is proportional to the sensed voltage signal developed across the second resistance, and φ is a phase angle between the sensed voltage signals developed across the first and second resistances, respectively.

15. The method of claim 11, wherein the control signal is useable to adjust one of the inductive circuit and the capacitive circuit.

16. The method of claim 11, wherein the control signal is useable to adjust each of the inductive circuit and the capacitive circuit.

17. The method of claim 11, wherein producing the control signal comprises performing a multiplier or comparison operation on the sensed signals.

18. The method of claim 17, wherein producing the control signal comprises low-pass filtering an output of the multiplier or comparison operation.

19. The method of claim 11, wherein the source device comprises an RFID reader and the untethered device comprises an RFID tag.

20. The method of claim 11, wherein the source device comprises a digitizer and the untethered device comprises a stylus.