Also at IEEE-IAS Annual Meeting, San Diego, October 1989

Significance: 

Part 4 – Propagation and coupling of surges 

In the early 80’s, considerable interest arose in developing an IEC standard test method to assess the immunity of 

electronic equipment against “fast transients” such as those that can be produced by contact bouncing in power circuits. 

 Such transients could propagate from their source – the bouncing contact – to the power port of equipment, or be 

coupled by proximity into control cables connected to the equipment of interest. The issue was not so much the 

potential for damaging the equipment – as in surge occurrences -- but rather the possibility of disturbing proper 

operation of the equipment. 

Motivated by some lingering skepticism among US industry on the necessity to require these tests across-the-board for 

industrial equipment, measurements, augmented by theoretical numerical simulations were conducted on representative 

power wiring configurations to help determine how distance of propagation will reduce the threat. The findings, showing 

good agreement between theory and experiments, validated the expectation, that is, substantial attenuation occurs when 

these fast transients travel more than a few tens of meters away from their source. See also file “Propag EFT1 1987” in 

this Part 4 for earlier tests on different cable configurations and numerical modeling technique. 

Filename: Propag EFT2 1990 
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS. VOL. 26. NO. I. JANUARYIFEBRUARY 1990 151 

Electrical Fast-Transient Tests : Applications and 

Limitations 

Abstract-According to a new standard of the International Electrotechnical Commission (IEC), a fast-transient test must be applied to 

the connecting cables of electronic equipment. The purpose of the test 

is to demonstrate equipment immunity to fast transients resulting from 

switching. Tests and simulations of the propagation and attenuation of 

these fast transients in typical connecting cables are described, placing 

the IEC requirements in perspective. 

C ONTINUING technical progress in the performance of 

electronic control systems has been made possible by 

fast logic devices operating at low levels of voltage and 

power, with high component density and interactive peripherals. These undisputed advantages, however, can make the circuits increasingly sensitive to external interfering overvoltages 

resulting in amplitude-related hardware failures or rate-ofchange-related system upsets. To guard against such effects, 

immunity to such failure must be demonstrated by realistic 

tests which validate the design rules and installation practices 

of the equipment. However, a balance must be found between 

overdesign and cost limitations. 

In contrast with traditional surge testing aimed at validating 

insulation levels or the energy-handling capability of surge 

protection devices, recent proposals advocate fast-transient 

tests demonstrating interference immunity. The new test waveforms represent electrostatic discharge effects or reignitions 

occurring during switching sequences. The basis for such proposals is an increasing awareness of the significance of fast 

transients. This new awareness results from two factors: 1) 

increases in the bandwidth and writing speed of oscilloscopes 

which makes possible the detailed observation and measurement of these phenomena, and 2) field failures of equipment 

which pass traditional tests but fail when subjected to the new 

fast-transient tests. 

The rationale for requiring the new tests is based on the 

assumption that interfering transients generated by power circuit switching will couple into adjacent power or signal lines. 

These interfering transients will then propagate toward susPaper PID 89-09, approved by the Petroleum and Chemical Industry Committee of the IEEE Industry Applications Society for presentation at the 

1989 Petroleum and Chemical Industry Committee Technical Conference, 

San Diego, CA, September 11-13, and at the 1989 IEEE Industry Applications Society Annual Meeting, San Diego, CA, October 1-5. Manuscript 

released for publication February 22, 1989. 

The authors are with the Electricity Division, National Institute of Standards and Technology, Building 220, Room B-344, U.S. Department of Commerce, Gaithersburg, MD 20899. 

IEEE Log Number 8931459. 

TIME 

(MICROSECONDS) 

Typical transient produced by contact opening. 1: voltage recovery 

of circuit (kHz frequency); 2: reignition (ns collapse); 3: complete recovery 

(ps duration). 

ceptible equipment. The fast transients contain high-frequency 

components that are expected intuitively to suffer greater attenuation than low-frequency components as they propagate 

in the lines. Verifying and quantifying this intuitive expectation provides a perspective on the severity of the situation 

and helps define realistic test requirements. To that end, this 

paper describes specific measurements conducted for typical 

low-voltage power line configurations. An attenuation model 

is proposed which provides a tool for understanding the significance of the line parameters and extends the usefulness of 

the results to general cases. 

THE IEC ELECTRICAL FAST TRANSIENT TEST 

The Technical Committee TC65 of the International Electrotechnical Commission (IEC) has completed a document [l] 

requiring immunity of industrial process control equipment to 

fast transients be demonstrated. According to that document, 

fast transients must be applied to the incoming power lines as 

well as to the inputloutput data lines of this equipment. 

The origin of fast transients is the well-known behavior of some air contactors when they interrupt an 

inductive-capacitive circuit: as the contacts part, the arc drawn 

between the contacts becomes unstable and the current is then 

interrupted, only to be re-established promptly by reignition 

across the gap. A race occurs between the dielectric strength 

of the increasing gap, and the voltage recovery of the circuit. 

Consequently, a sequence of clearing and reignition ensues, 

with an eventual full clearing of the circuit. As shown in 

Fig. 1, each of the clearings produces the beginning of an 

oscillation at relatively low frequency depending on the local 

circuit parameters. As the voltage across the gap increases 

at a rate corresponding to this frequency, reignition occurs if 

the gap withstand voltage is exceeded. The waveform of each 

0093-999419010100-015 1$01.00 O 1990 IEEE 
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS. VOL. 26, NO. I. JANUARYIFEBRUARY 1990 

VOLTAGE 

t 

VOLTRGE 7 Pulse 

I / Spacing of single pulses depends 

on the test voltage level 

VOLTAGE 7 Burst 

1 Burst length 1 

1- Burst period - 300 ms -4 

Fig. 2. Fast-transients specified by IEC 801-4. Top: waveshape of single 

pulse for 50-0 load. 

sequence is then a series of pulses, each with a relatively slow 

voltage rise followed by collapse in a few nanoseconds. 

Thus the circuit behaves as a generator of transient pulse 

bursts. Each pulse has a slow rise ending in an abrupt collapse. 

Because the pulses occur in bursts, these single pulses have 

the same effect as pulses with fast rise time and slower decay. 

For standardization purposes, the IEC document specifies an 

electrical fast transient (EFT) 5150-ns waveform as shown in 

Fig. 2, where the rise time is 5 ns and the duration (full width 

at half maximum) is 50 ns. The repetition rate of the pulses 

within a burst is not specified. However, the duration of the 

burst and the interval between bursts are specified. 

Three limitations exist in the severity of the situation, resulting from the fast transient nature of the test and the conditions 

of application. 

1) As described in the IEC test procedure, the test transients are coupled to the equipment under test (EUT) by a 

capacitive "clamp" through which the power or data cable is 

inserted. This clamp is a mechanical test fixture consisting of 

two hinged metallic plates designed to provide a fixed and repeatable capacitance between the output of the pulse generator 

and the input cables. Discrete capacitors can also provide the 

coupling. This coupling method produces a capacitive divider: 

the high side of the divider is the coupling capacitor, and the 

low side is the internal capacitance to ground of the EUT (Fig. 

3). This effect, therefore, is influenced by the design of the 

DATA OR 

POWER 

CABLE 

COUPLING CLAflP 

EUT 

A I 

~n 

I 

I 

I 

I 

T -Y I I I 

GROUND PLANE 

Fig. 3. Coupling electrical fast transient (EFT) pulses into equipment cables. 

equipment and becomes a design parameter controlled by each 

manufacturer. 

2) The freedom left to those who perform the test on setting the repetition rate of the pulses leaves some ambiguity 

on the total number of the pulses applied to the EUT. The 

test presumably aims at identifying an interference with the 

logic of the circuit. That interference is likely to occur during 

critical state transitions of the semiconductor circuitry of the 

equipment, malung timing and coincidence very significant. 

Advocates of electrostatic discharge tests make a case that up 

to 10000 pulses should be applied at random to produce a 

high degree of confidence that the issue of pulse timing has 

been addressed 121. 

3) In an actual installation, attenuation between the point of 

coupling and the victim equipment will reduce the severity of 

the impinging transient. The line propagation characteristics 

reduce the peak amplitude and increase the rise time of the 

pulse as it travels down the line. This effect, therefore, is 

influenced by the line regardless of this equipment design. 

Although our measurements and modeling address this third 

limitation, the two other limitations should also be kept in 

mind. 

Two line configurations were selected as typical of lowvoltage power lines: a three-conductor line in a steel conduit, 

and a three-conductor cable with a nonmetallic jacket. The 

steel conduit line is easy to set up in the laboratory by arranging conduit sections in a zig-zag pattern. This arrangement 

allows placing the start and the end of the line next to each 

other to minimize the distance between the oscilloscope used 

for measuring the signals at either end of the line. At one end 

of the line, called the "sending end," single pulses with the 

IEC EFT waveform were injected. At the other end, called 

the "receiving end," the arriving pulses were recorded for 

various line terminating impedance configurations. The same 

oscilloscope was used to measure the waveforms at the two 

ends of the line. Thus the voltage ratios of the waveforms do 

not depend on the absolute accuracy of the instrumentation. In 

addition, the differential connection of the oscilloscope probes 

allows measurement without introducing an additional ground 

in the line, which is already grounded by its connection to the 

pulse generator. This arrangement also removes any question 

of variations of the stray capacitances to ground, provided that 
MARTZLOFF AND LEEDY: ELECTRICAL FAST TRANS~ENT TESTS 153 

the instruments stay in the same position for the two measure- loO1 /\ 1 

ments, at both the sending and receiving end. The experi- 90 

MEASURED INPUT PULSE mental details of the measurements are discussed in , 8o MODEL - WIRES NEAR CENTER OF CONDUIT 

Appendix I. 2 

a 70 

Because the conduit contains the electromagnetic fields as- 5 

4 60 

sociated with the propagating pulse, there is no need to be : 

concerned with coupling between adjacent sections of the line a 

in the zigzag, nor with losses by radiation from the line act- a * 

ing as an antenna. For the nonmetallic jacketed line, however, $ so 

these effects must be addressed. - $ 20 

To investigate the effect of radiation losses from the non- ' ODEL - WIRES NEAR WALL OF CONDUIT 

metallic jacketed line, three experiments were performed outdoors. First, measurements were made with the line arranged 0 

0 10 20 30 40 60 70 80 90 

in a wide loop, elevated from ground and away from metallic TIME (NANOSECONDS) 

objects. TO assess the importance of the proximity of the line Fig. 4. Composite plot of measured and modeled pulses, as percent of 

to ground, the line was later dropped to ground level and a amplitude, for wires enclosed in metal conduit. 

second set of measurements was made. Last, a coil of identical cable was also subjected to the same measurements. The 

observed difference of the three sets of measurements was 

negligible for practical purposes. 

For the case of the conduit-enclosed line, a previous paper 

reports in detail the experimental results and theoretical expectations from a model solely based on the geometry of the 

line [3]. As a convenience for the readers and for comparison 

with the new results on nonmetallic jacket line, the results of 

the first configuration are summarized here. For the nonmetallic line, we used a new model based on measurable electrical 

parameters of the line, rather than the geometry of the line. 

Appendix I gives details on instrumentation considerations; 

Appendix I1 gives details on the line propagation model. 

Steel Conduit Line 

A representative low-voltage indoor single-phase power line 

was installed in the laboratory, with three conductors in a steel 

conduit. The line length was 63 m (20 thin-wall conduit sections, each the standard 10-ft length), using typical industrial 

components such as wire nuts at the splices in the wire. Three 

insulated (600 V) copper conductors of number 14 AWG 

( 1.6 mm diameter), pulled in the 20-mm diameter conduit ( 112 

EMT), represent the phase, neutral, and grounding conductors. At the sending end, the neutral conductor was bonded to 

the grounding conductor and to the conduit. 

At the receiving end, the neutral conductor was not bonded 

to the grounding conductor, nor to the conduit. This type 

of connection is typical U.S. practice for a branch circuit 

originating at a service entrance. 

For modeling purposes, it is much easier to assume that 

the line is uniform. In practice, the location of the conductors 

vary within the conduit due to sag, bends, or crossed wires. 

Therefore, we considered two limiting cases: 1) the wires are 

located at the center of the conduit, separated from each other 

by the thickness of the wire insulation, and 2) the wires are 

located against the wall of the conduit. In the numerical analysis, a model parameter k accounted for these two positions. 

A detailed description of the analysis is given in [3]. 

Fig. 4 shows the composite results of the measurements and 

modeling for the IEC EFT pulse arriving at the receiving end. 

The results are expressed as a percentage of the pulse applied 

Fig. 5. Oscillograms of measured pulses, as percent of input amplitude 

(0 m), for various lengths of wires enclosed in metal conduit (25, 38, and 

63 m). 

at the sending end. All the measurements were made with a 

injected pulse of 2.5 or 5 V, to avoid the problems of electromagnetic interference or safety associated with the levels 

of 500-4000 V specified by the IEC standard. The linearity 

of the circuits allows this simplification. The pulse, measured 

at the receiving end, is 50 percent of the sending-end pulse 

and lies between the two limiting cases of the model. Qualitatively, the model pulses look very much like the measured 

pulses. From the position of the measured pulse with respect 

to the two limiting cases, we can then assign a value to the 

model parameter k such that attenuation for the 63-m line is 

accurately matched. Given this empirically determined value, 

the model will show the effect of varying the line length. 

Measurements for several line lengths (Fig. 5) show 

receiving-end peak amplitudes of 76 and 65 percent of the 

sending-end peak amplitude for 25- and 38-m lines, respectively. Corresponding peak amplitudes predicted by the model 

are 75 and 67 percent (Fig. 6). For the greater line lengths, 

where measurements would be cumbersome, the model can 

now readily provide the answers. Fig. 6 includes examples 

for 120- and 240-m lengths, showing the continued attenuation and smearing of the pulse as the line length increases. 

Nonmetallic Jacketed Line 

For this type of line, the insulation holds the conductors in 

a fixed and uniform geometry relative to each other, in contrast with the random location of the conductors pulled in a 

conduit. It is possible to determine, by a simple set of measurements or calculations, the parameters of the transmission 
154 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS. VOL 26. NO I. JANUARYIFEBRUARY 1990 

0 10 30 50 70 90 110 130 150 

TIME INANOSECONDSI 

Fig. 6. Model prediction, as percent of input amplitude, for various lengths 

of wires enclosed in metal conduit. 

line: the series inductance L; the shunt capacitance C; and the 

series resistance R. The shunt admittance, normally included 

in the model of a transmission line, is negligible in a wellinsulated power line. For the high-frequency components of 

the fast-transient pulse, skin effect influences the effective series impedance. The model includes an approximation of this 

effect. 

The 75-m line (250 ft) consisted of a neoprene-jacketed cord 

with three number 14 AWG (1.6-mm diameter) neopreneinsulated copper conductors. The line was suspended 2 m 

above ground. The line was arranged in a triangular loop 

with the sending end and the receiving end at one corner of 

the triangle. In this manner, either end could be connected 

to the oscilloscope. As described in Appendix I, a constant 

impedance load was maintained at the sending end by using a 

dummy load when switching the oscilloscope from the sending 

end to the receiving end. In all the measurements, two of the 

conductors were bonded at the sending end, representing the 

neutral and the grounding conductors. At the receiving end, 

the arriving pulse was recorded for various terminating load 

combinations. Fig. 7 shows an EFT signal with 2.5 V peak 

and 5150-ns waveform injected at the sending end, arriving as 

a 3011 10-ns signal at the end of the line. This peak amplitude 

of 48 percent is very close to the 50-percent value found for 

the steel conduit. The small difference between the two values 

is due to the oscillogram reading uncertainty rather than to 

a significant difference between the characteristics of the two 

lines. The question raised in the introduction, concerning radiation losses, is answered in the negative-or at least in the 

form, "not discernable at this level of significance." 

Need to Consider Loading at Receiving End 

Classical transmission line theory points out that reflections 

occur at the receiving end of the line when the terminating 

impedance differs from the characteristic impedance of the 

cable: 1) doubling the arriving pulse for an open end (and thus 

reflecting a pulse of the same polarity); 2) maintaining zero 

for a shorted end (and thus reflecting a pulse of reverse polarity); and 3) maintaining the level unchanged for a terminating 

(b) 

Fig. 7. Pulses for nonmetallic jacketed cable. (a) Sending-end pulse. 

Vertical: 1 Vldiv, sweep: 20 nsldiv. Signal parameters: amplitude = 5 V 

(100 percent), rise time = 5 ns, FWHM = 50 ns. (b) Receiving-end pulse. 

Vertical: 0.5 Vldiv, sweep: 20 nsldiv. Signal parameters: amplitude = 2.4 

V (48 percent), rise time: 30 ns, FWHM: 110 ns. 

impedance that matches the line characteristic impedance (and 

thus having no reflection). 

Measurements made with "slow" 1.2150-ps impulses have 

shown that, while correct, this analysis does not produce useful results if applied to line lengths that do not contain the 

whole impulse front [4]. For a 1.2-ps front in a line where 

the propagation velocity is approximately 213 the speed of 

light in vacuum (300 mlps), this lower limit on line length 

is approximately 200 m. For the 5150-ns IEC EFT pulse, the 

63-m line produces at the receiving end a 30-ns front time. 

According to the model, a 240-m line would produce a 60-ns 

front time (Fig. 6). Clearly, then, for this short an impulse, 

the receiving-end terminating impedance effects must be considered according to classical transmission line theory. 

Because the measurement system has an impedance of 

100 R while the characteristic impedances of the lines investigated are approximately 60 R, the case of an open-ended line 

cannot be measured. In actual installations and in the absence 

of a low-impedance instrument, the pulse would have twice 

the amplitude shown in Fig. 7. An open end is the limiting 

case for a very light load such as a control circuit during the 

off cycle of the power circuit. Adding even a small admittance 

at the end of the line substantially reduces the doubling effect. 

Common-Mode and Normal-Mode Coupling 

The method specified by IEC 801-4 for coupling the test 

transients to the power or the data cables introduces them as 

a common-mode signal in the cable (Fig. 3). This situation 

is not explicitly pointed out in the IEC document, but it must 

be clearly understood. Another standard advocating fast transients [5] specifies the application of the test transients in any 
MARTZLOFF AND LEEDY: ELECTRICAL FAST TRANSIENT TESTS 155 

Fig . 8. 

small 

Superimposed oscillograms of pulses at receiving-end, showing 

difference between overhead line (left) and coiled line (right). 

possible combination of the conductors for all input and output ports. While space and scope limitations for this paper 

prevent a comprehensive discussion of the subject, alerting 

readers to the issue is worthwhile. 

The IEC EFT test focuses on installed equipment rather 

than on stand-alone equipment. The coupling clamp method 

for injecting the transients into the circuits is a very convenient procedure when cables are accessible. Practitioners of 

this test also report useful results by coupling the transients 

with foil wrapped around installed bundles of cables. The document provides detailed descriptions of the configuration for 

a laboratory test in an effort to ensure reproducible results. 

However, propagation, coupling, and attenuation effects associated with the capacitive divider nature of the test application 

will be strongly affected by small changes in the test circuit 

geometry, making the test difficult to describe and reproduce 

in absolute terms. 

These remarks should not be construed as a criticism of the 

test. Anecdotal field experience shows that equipment with 

demonstrated immunity to the IEC 801-4 EFT procedure is 

less likely to be disturbed than equipment not meeting this 

requirement. The addition of the new test procedure has contributed to improved reliability. Experience is the ultimate validation of the usefulness of any performance standard. 

No Practical Need for Outdoor Tests 

After completing the tests on the suspended aboveground 

line, the line was lowered to ground (dry grass) level, and one 

set of measurements was made under that condition. The same 

measurements were also made on a control coil of identical cable, left in its original packing carton. These two control measurements show detectable but insignificant differences 

(Fig. 8) in the pulse arriving at the receiving end. Thus concern over line geometry and coiled versus "free space" configurations, while technically correct, is a negligible factor in 

practical applications. Therefore, future experimenters need 

not go outdoors to perform measurements, even at the higher 

frequencies since the coiled versus free space configurations 

appear to produce similar results. This finding extends the 

conclusion previously reached for 100-kHz ring waves 161. 

Line Model 

An electrical circuit analysis model was devised to investigate the changes in the voltage amplitude and waveform shape 

OUTER NEOPRENE JACKn -\ 

INNER NEOPRENE 

INSULATION 

STRANDED CC 

CONDUCTOR 

'ER - 

FILLER MATERIAL - FA?(': 

Fig. 9. Section of three-conductor, nonmetallic jacketed cable. 

of the 5150-11s EFT pulse as it propagates along a length of 

typical low-voltage power line. The low-voltage cable consists of three number 14 (1.6-mm diameter) conductors, with 

individual neoprene insulation and covered with a neoprene 

jacket. Fig. 9 shows a cross section of this cable. 

A 63-m length of cable was simulated as 63 1-m lengths 

connected in series (Fig. 10). Each 1-m length was represented 

by the equivalent twoport subcircuit shown in Fig. 11. The 

subcircuit values represent the series and parallel elements of 

the two-conductor line, to which a capacitance from each wire 

to a hypothetical ground plane was added. Numerical values 

were assigned to these parameters as described in Appendix TT 

11. 

Electrical circuit analysis simulations were performed using 

the SABER software system' which allows the 1-m subcircuit 

of Fig. 11 to be described only once as a "template." The 

software template is then replicated as many times as necessary 

to represent the transmission line. To investigate the effects of 

a change of a modeling parameter, such as a resistance or 

capacitance, the parameter need be changed only once in the 

data set. The 5150-11s IEC EFT pulse was digitized from an 

oscillogram recorded during the experimental measurements 

so that a piecewise linear voltage source could be constructed 

for the circuit simulation. 

Modeling Results 

Fig. 12 shows the comparison of the observed (solid line) 

and the simulated (dashed line) waveforms at the end of 

63 m of cable in response to the 5150-ns IEC EFT pulse. 

The pulse simulated by the simple model described demonstrates the same general waveshape, rate of voltage rise, and 

amplitude as the observed pulse. The simulated pulse, however, has a shorter time duration than the observed pulse. In 

addition, the propagation velocity of the pulse, as predicted 

by the model, is slightly greater than the propagation velocity 

measured in the cable. 

The discrepancies between the simulated and the observed 

waveforms and the velocities may be explained by several limitations of the model. First, the model did not consider the skin 

effect of the conductors. The skin effect would raise the resistance and inductance of the conductors for the high-frequency 

components of the waveform. Secondly, the cable consisted of 

'SABER is a product of Analogy, Inc., Beaverton, OR. 
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS. VOL. 26. NO. I, JANUARYIFEBRUARY 1990 

TERMINATION SOURCE 

- c 63 SECTIONS REPRESENTING 63m CABLE - 

Fig. 10. Building line with repeated sections. 

A pragmatic conclusion on the quantification of the EFT 

pulse propagation is that equipment likely to be installed close 

to sources of such disturbances would benefit from immunity 

demonstration; equipment intended to be located away from 

such sources, however, might not be required to be subjected 

to a complete set of EFT tests. 

Fig. 11. Circuit parameters for individual sections. 

Fig. 12. Comparison behveen measured and modeled pulses at receiving 

end of nonmetallic jacketed line. Solid line: measured data; dashed line: 

predicted data. 

three conductors of which only two were modeled. The third 

conductor was connected to ground at the pulse generator end 

of the cable and left open at the receiving end of the cable. 

The presence of this third conductor would change the propagation characteristics of the cable slightly. Finally, a simple 

loss model (a series RC network) has been assumed for the capacitance associated with the neoprene dielectric of the cable. 

The response at high frequencies of organic dielectrics may 

be considerably more complicated than this model reflects. 

The two line configurations and their models yield similar 

results for measurements and for computations. Some difference might be expected between the quasi-coaxial configuration of a conduit-enclosed line and the open-line configuration 

with unshielded conductors of a nonmetallic jacket. 

The effects of propagation of an EFT pulse in the lines are 

a reduction in the amplitude, a decrease in the steepness of 

the front, and an increase in the duration of the pulse. The 

first two effects reduce the severity of a pulse arriving at the 

victim equipment; the third effect is less significant for victim 

equipment sensitive to rate-of-change disturbances. 

The two models provide the tools for analysis of other configurations. The choice of the geometry model or the electrical parameter model will depend on the available software 

and preferences of the investigator. 

P. F. Wilson (former staff member of the National Bureau 

of Standards, Boulder, CO) developed the model [3] for the 

conduit line, which is only summarized here. The nonmetallic jacket measurements were funded in part by a consortium 

of private sector companies coordinated by Building Industry 

Consulting Service International of Tampa, FL. 

The voltages were recorded with a 1-GHz bandwidth oscilloscope and two 50-0 input vertical preamplifiers. The mainframe of the oscilloscope performed the differential combination of the two signals. In this configuration, the bandwidth of 

the system is only 500 MHz, compared to the 1-GHz capability of the single-ended measurements with one preamplifier. 

This reduction in performance is an acceptable trade-off to 

obtain the differential connection of the probes. In the experiments, the conduit, oscilloscope chassis, and low-side output 

of the generator are bonded together and hence ensure minimum noise coupling. Where high voltages are involved, this 

configuration also provides improved safety. This was not a 

concern here as the measurement on this linear system were 

conducted with signals in the 5-V range. A check on the common mode rejection of the system indicated less than fivepercent residual signal, an insignificant effect compared to 

the attenuation levels measured. 

The preamplifiers have a 50-R input impedance, so that the 

effective impedance of the differentially connected probe is 

100 R. At the sending end, the voltages of the line and neutral 

conductors with respect to chassis ground were each fed to 

one of the preamplifiers (Fig. 13). Thus the pulse delivered 

by the generator impinges on an impedance consisting of the 

two amplifiers in series, in parallel with the line. To make 

measurements at the receiving end, the same oscilloscope was 

used with its differential probes. However, the pulse arriving 

at the receiving end would experience a reflection due to any 

impedance mismatch. To obtain an impedance match, timedomain reflectometry was used to trim the terminating load 

until a minimal reflection occurred. 

Good matching occurred with a 75f2 termination for the 

steel conduit line (300-R trim in parallel with the 100-9 

probes). For the nonmetallic-jacketed cable, a 50-R termination gave best result (100-R trim and 100-f2 probes in parallel). For the steel conduit line, a dummy load was used, made 
MARTZLOFF AND LEEDY: ELECTRICAL FAST TRANSIENT TESTS 

MATCHING PULSE 

GENERATOR RESISTOR 7 

OSCILLOSCOPE 

PULSE 

GENERATOR ------ ---------- 

CATHODE RAY 

osc'LLoscopE (y] 

1 I 

& 

(b) 

Fig. 13. Configuration for constant-load measurements with dummy load 

and single oscilloscope. (a) Measurement at sending end. (b) Measurement 

at receiving end. 

SPECTRUM ANALYZER INPUT 

6.23 TEST LINE 

P 

Fig. 14 

COAXIAL-TO-BINSING 

POST ADAPTER 

COAXIAL CABLE 

- 40 MHz 

TRACKING GENERATOR OUTPUT 

Configuration for determining velocity of propagation in nonmetallic jacketed line. 

of two 50-R coaxial terminators to provide the same matching impedance as that of the oscilloscope probes, as shown 

in Fig. 13. For the nonmetallic-jacketed cable measurements, 

two preamplifiers of the same type as the measuring preamplifiers, in a compatible mainframe, were used as the dummy 

load. In this manner, when recording the receiving-end waveforms with the measurement oscilloscope, the dummy load 

connected in parallel with the line at the sending end maintained a constant impedance load on the pulse generator. 

The propagation velocity in the cable was measured as reported previously: a 40-MHz spectrum analyzer, with a tracking generator output, was used to determine the frequencies 

at which the input impedance of a length of cable is minimum 

[6]. Fig. 14 shows schematically the test line, the spectrum 

analyzer, and a resistive attenuator to improve the decoupling 

of the tracking generator output from the test line. 

Fig. 15 shows the spectrum for a 6.23-m length of opencircuited test line. Voltage minima occur at 5.64, 19.4, 

and 30.4 MHz, corresponding to wavelengths of hl4, 3M4, 

10 2 0 3 0 40 

FREQUENCY (MHz) 

Fig. 15. Spectrum of 6.23-111 length of test line. 
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS. VOL. 26. NO. I. JANUARYIFEBRUARY 1990 

Fig. 16. Circuit parameters for line sections. 

and 5hl4, respectively. The average propagation velocity, the the capacitance between two parallel cylinders representing 

product of h and frequency, is 155 mips. the conductor-to-conductor capacitance may be written as [8] 

The series inductance of the 1-m section L, was calculated 

using the formula for the inductance of two parallel conductors in close proximity to each other, with current in opposite 

directions in each conductor [7]. For such a situation, the effective inductance L, is given in engineering handbooks as 

where 

L, effective inductance (nH), 

I length of the cable (in), 

d distance between the conductors (in), 

r radius of the conductors (in), 

p relative magnetic permeability (copper = 1.0), 

6 skin depth factor. 

where C, is the capacitance per unit length (pF11000 ft) and 

E, is the relative dielectric constant of the material filling the 

space between the two conductors. The equation for the capacitance between a wire and a ground plane is similar; this 

capacitance is 

where C, is the capacitance per unit length (pFi1000 ft) and 

H is the distance between the conductor and the ground plane. 

The resistance of the conductor is represented by resistance 

R, in series with each inductance L,. A resistance, R, is 

in series with the capacitance C, to simulate the losses encountered in the dielectric surrounding the two conductors. 

One additional resistance R, simulates the resistance associated with the conductor-to-conductor leakage to complete the 

model. 

where 

K = 3168, a conversion constant, 

p volume resistivity (0-in) ( = 0.68 . lop6 for copper), 

f frequency (Hz). 

For a low-frequency approximation of the response of the cable to a 5150-ns pulse, the skin effect term can be neglected. 

There are two types of capacitance associated with the cable (Fig. 16). The first is associated with the conductor-toconductor capacitance, C, . The other type of capacitance consists of equd valued capacitors C, I and Cg2, which represent 

the two conductor-to-ground capacitances. The equation of 

sumes E, = 3 for neoprene dielectric; 

C, 17 pF, the conductor-to-ground capacitance; 

R, 0.003 R, the series resistance of conductor (obtained 

from wire tables); 

R, 10 R, the estimated resistance in series with C,; 

R, 10 MR, the estimated leakage resistance. 

The velocity of propagation predicted by the model is approximately 170 mlps for the 5150-ns pulse traversing the 

neoprene-jacketed cable. In contrast, the higher velocities of 

propagation would be expected in the line enclosed by the 

steel conduit where the conductors have only a thin insulation 

surrounded by air. Measurements of the velocity of propagation confirm the difference between the two lines: 210 mlps 

for the air-surrounded conductors in the conduit and 155 mlps 

for the neoprene covered conductors. 
MARTZLOFF AND LEEDY: ELECTRICAL FAST TRANSIENT TESTS 159 

REFERENCES tions, and interference mitigation. In addition to his research work on surge 

,qectrical ,cast ~~~~~i~~~~~~i~~~~~IEC publication 801-4, propagation and mitigation, he is contributing to the development of stanGeneva, Switzerland, 1988. dards on surge environment and surge suppression within IEEE, ANSI, and 

R. B. Bush, "Statistical considerations of electrostatic discharge evalu. IEC by documenting his measurements, as in the present paper, by resenti in^ 

ations," in Pm. Int. Zurich Symp. Electromagnetic Compatibility, and by new standards. 

1987. 

F. D. Martzloff and P. F. Wilson, "Fast transient tests: Trivial or terminal pursuit?" in Proc. Znt. Zurich Symp. Electromagnetic Compatibility, 1987. 

F. D. Martzloff, "The propagation and attenuation of surge voltages 

and surge currents in low-voltage ac power circuits," IEEE Trans. 

Power App. Syst., vol. PAS-102, no. 5, May 1983. 

Standard Surge Withstand Capbility (SWC) Tests for Protective 

Relays and Relay Systems, ANSIIIEEE C37.90.1, 1989. 

F. D. Martzloff and H. A. Gauper, "Surge and high-frequency propagation in industrial power lines," IEEE Trans. Ind. Appl., vol. IA-22, 

no. ~-. . 4. ,.-., JUIV/AIIP. ~ 1986. 

P. M. Rostek, "Avoid wiring-inductance problems," Electron. Design, Dec. 6, 1974. 

H. Pender and W. A. Del Mar, Electrical Engineers' Handbook. 

New York: Wiley (Engineering Handbook Series), 1949, p. 14-33. 

Fkanqois D. Martzloff (M'56-SM'80-F'83) completed his undergraduate studies in France and received the M.S.E.E. degree from the Georgia Institute of Technology, Atlanta, and the M.S.I.A. degree from Union College, Schenectady, NY. 

After a long career at General Electric, he joined 

the staff of the National Bureau of Standards (now 

renamed National Institute of Standards and Technology) to work on conducted electromagnetic interference issues. His early experience covered highvoltage fuses, high-voltage bushings, and electronic 

power conversion; the latter marked a turning point to issues of overvoltage 

effects on semiconductors, surge suppression, metal oxide varistor applicaThomas F. Leedy (S'64-M'65) received the B.S. 

degree in physics from the American University, 

Washington, DC, and the M.S. degree in electrical 

engineering and the E.Ad. degree in engineering 

administration from George Washington University, 

Washington, DC, in 1965, 1972, and 1984, respectively. 

From 1965 to 1980 he did research in microelectronics at the National Bureau of Standards where 

he developed and validated measurement methods 

for determining the effects of transient high-energy 

radiation on integrated circuits and for evaluating materials and processes of 

microelectronic device fabrication. Since 1980, he has been a member of the 

technical staff of the Electronic Instrumentation and Metrology Group of the 

Electrosystems Division where he has participated in the research program 

directed toward assuring the accuracy of automatic test equipment. 

Mr. Leedy has served on numerous standards committees in microelectronics, computers, radiation effects, and electronic instrumentation for IEEE, 

ASTM, and the U.S. Department of Commerce International Trade Administration.