# Patent

## Title
Programmable gain amplifier (PGA) with AGC in receiver section

## Abstract

A method and apparatus for dynamically controlling a programmable gain amplifier (PGA) to provide a plurality of gain steps thereby providing automatic gain control (AGC) in a receiver intermediate frequency (IF) stage comprising a succession of operational amplifiers wherein at least one is for providing fine gain control and wherein the gain of each fine gain amplifier is controlled by the resistance ratios of a plurality of selectively biased MOSFETs. In one embodiment of the invention, three coarse amplifiers are provided, each having a gain of either 0 dB or 12 dB based on the value of a two-state signal provided to each amplifier gain control input. A single fine gain amplifier has a gain of 0 dB, 3 dB, 6 dB or 9 dB based on the binary value of the two-bit signal provided to the amplifier.

## Background
CROSS-REFERENCE TO RELATED APPLICATION
This application is a continuation of and claims priority to U.S. Utility application Ser. No. 10/138,602, filed May 3, 2002, now U.S. Pat. No. 6,862,438 having a title of “Programmable Gain Amplifier (PGA) with AGC in Receiver Section” which application is incorporated herein by reference in its entirety for all purposes.
This application claims priority to and incorporates by reference U.S. Provisional Application entitled, “Method and Apparatus for a Radio Transceiver”, having a Ser. No. 60/367,904 and a filing date of Mar. 25, 2002.


BACKGROUND
1. Field of the Invention
This invention relates generally to wireless communications and, more particularly, to the operation of a Radio Frequency (RF) transceiver within a component of a wireless communication system.
2. Description of the Related Art
The structure and operation of wireless communication systems are generally known. Examples of such wireless communication systems include cellular systems and wireless local area networks, among others. Equipment that is deployed in these communication systems is typically built to support standardized operations, i.e., operating standards. These operating standards prescribe particular carrier frequencies, modulation types, baud rates, physical layer frame structures, MAC layer operations, link layer operations, etc. By complying with these operating standards, equipment interoperability is achieved.
In a cellular system, a regulatory body typically licenses a frequency spectrum for a corresponding geographic area (service area) that is used by a licensed system operator to provide wireless service within the service area. Based upon the licensed spectrum and the operating standards employed for the service area, the system operator deploys a plurality of carrier frequencies (channels) within the frequency spectrum that support the subscriber units within the service area. Typically, these channels are equally spaced across the licensed spectrum. The separation between adjacent carriers is defined by the operating standards and is selected to maximize the capacity supported within the licensed spectrum without excessive interference. In most cases, severe limitations are placed upon the amount of adjacent channel interference that may be caused by transmissions on a particular channel.
In cellular systems, a plurality of base stations is distributed across the service area. Each base station services wireless communications within a respective cell. Each cell may be further subdivided into a plurality of sectors. In many cellular systems, e.g., Global System for Mobile Communications (GSM) cellular systems, each base station supports forward link communications (from the base station to subscriber units) on a first set of carrier frequencies, and reverse link communications (from subscriber units to the base station) on a second set of carrier frequencies. The first set and second set of carrier frequencies supported by the base station are a subset of all of the carriers within the licensed frequency spectrum. In most, if not all, cellular systems, carrier frequencies are reused so that interference between base stations using the same carrier frequencies is minimized and system capacity is increased. Typically, base stations using the same carrier frequencies are geographically separated so that minimal interference results.
Both base stations and subscriber units include RF transceivers. Radio frequency transceivers service the wireless links between the base stations and subscriber units. The RF transmitter receives a baseband signal from a baseband processor, converts the baseband signal to an RF signal, and couples the RF signal to an antenna for transmission. In most RF transmitters, because of well-known limitations, the baseband signal is first converted to an Intermediate Frequency (IF) signal and then the IF signal is converted to the RF signal. Similarly, the RF receiver receives an RF signal, down converts the RF signal to an IF signal and then converts the IF signal to a baseband signal. In other systems, the received RF signal is converted directly to a baseband signal.
In the initial signal processing stages of an RF receiver, the received RF signal is converted to baseband through one or more steps. Initially, the received RF is mixed with a local oscillator (LO) to down convert the carrier frequency of the RF. It is also common to utilize a low pass filter coupled to the output of the mixer to remove introduced interferences. One problem with using a mixer and low pass filter, however, is that both add a fixed amount of gain to the received signal, irrespective of the received signal strength.
This amplified signal is passed to the baseband processor, which is sensitive to excessive signal strength which can cause quantification errors in downstream analog-to-digital converters (ADCs) and saturation in the output devices. Previous designs have included a programmable gain amplifier (PGA) between the low pass filter output and the baseband processor input to limit the gain when the received signal strength is excessive. This approach, however, requires a method to detect the peak amplitude of the received signal then provide a control signal to the PGA that is proportional to the received signal amplitude.
Since a received signal is comprised of a differential I and Q signal, the traditional method of detecting the peak amplitude is to take the square root of the sum of I squared and Q squared [(I2+Q2)1/2]. The square root function is traditionally performed using a digital signal processor (DSP) in the digital domain. Generally, though, it is advantageous to perform such peak amplitude detection in the analog domain and to perform gain control based upon the detected peak values. There is a need in the art, therefore, for an analog circuit and a method to adjust the gain level of the PGA according to the determined peak amplitude of the amplified signal.
SUMMARY OF THE INVENTION
The present invention is directed to apparatus and methods of operation that are further described in the following Brief Description of the Drawings, the Detailed Description of the Invention, and the claims. Other features and advantages of the present invention will become apparent from the following detailed description of the invention made with reference to the accompanying drawings.



BRIEF DESCRIPTION OF THE DRAWINGS
These and other features, aspects and advantages of the present invention will be more fully understood when considered with respect to the following detailed description, appended claims and accompanying drawings wherein:
FIG. 1A is a system diagram illustrating a cellular system within which the present invention is deployed;
FIG. 1B is a block diagram generally illustrating the structure of a wireless device constructed according to the present invention;
FIG. 2 is a block diagram illustrating a subscriber unit constructed according to the present invention;
FIG. 3 is a functional schematic diagram of an amplifier system;
FIG. 4 is an amplifier formed to have programmable gain;
FIG. 5 is a functional schematic block diagram of an RF processing unit of a radio transceiver formed according to one embodiment of the present invention;
FIG. 6 is a functional schematic diagram illustrating one embodiment of the programmable gain amplifiers used in the present invention;
FIG. 7 is a functional schematic diagram of an amplifier that provides gain in 3 dB steps;
FIG. 8 is a functional block diagram that illustrates one embodiment of the present invention for controlling the programmable gain amplifiers for the I and Q modulated channels;
FIG. 9 is a functional schematic block diagram that illustrates an analog peak detector circuit formed according to one embodiment of the present invention; and
FIG. 10 is a functional schematic block diagram that illustrates one embodiment of the present invention.



DETAILED DESCRIPTION OF THE DRAWINGS
FIG. 1A is a system diagram illustrating a cellular system within which the present invention is deployed. The cellular system includes a plurality of base stations 102, 104, 106, 108, 110, and 112 that service wireless communications within respective cells, or sectors. The cellular system services wireless communications for a plurality of wireless subscriber units. These wireless subscriber units include wireless handsets 114, 118, 120, and 126, mobile computers 124 and 128, and desktop computers 116 and 122. During normal operations, each of these subscriber units communicates with one or more base stations during handoff among the base stations 102 through 112. Each of the subscriber units 114 through 128 and base stations 102 through 112 include RF circuitry constructed according to the present invention.
The RF circuitry formed according to the present invention may be formed to operate with any one of a number of different protocols and networks. For example, the network of FIG. 1A may be formed to be compatible with Bluetooth wireless technology that allows users to make effortless, wireless and instant connections between various communication devices such as notebook computers, desktop computers and mobile phones. Because Bluetooth systems use radio frequency transmissions to transfer both voice and data, the transmissions occur in real-time.
The Bluetooth specification provides for a sophisticated transmission mode that ensures protection from interference and provides security of the communication signals. According to most designs that implement the Bluetooth specifications, the Bluetooth radio is being built into a small microchip and is designed to operate in frequency bands that are globally available. This ensures communication compatibility on a worldwide basis. Additionally, the Bluetooth specification defines two power levels.
A first power level covers the shorter, personal area within a room and a second power level is designed for covering a medium range. For example, the second power level might be used to cover communications from one end of a building, such as a house, to the other. Software controls and identity coding are built into each microchip to ensure that only those units preset by the owners can communicate with each other. In general, it is advantageous to utilize low power transmissions and components that draw low amounts of power (especially for battery operated devices). The Bluetooth core protocols include Bluetooth-specific protocols that have been developed for Bluetooth systems. For example, the RFCOMM and TCS binary protocols have also been developed for Bluetooth but they are based on the ETSI TS 07.10 and the ITU-T recommendations Q.931 standards, respectively. Most Bluetooth devices require the Bluetooth core protocols, in addition to the Bluetooth radio, while the remaining protocols are only implemented when necessary.
The baseband and link control layers facilitate the physical operation of the Bluetooth transceiver and, more specifically, the physical RF link between Bluetooth units forming a network. As the Bluetooth standards provide for frequency-hopping in a spread spectrum environment in which packets are transmitted in continuously changing defined time slots on defined frequencies, the baseband and link control layer utilizes inquiry and paging procedures to synchronize the transmission of communication signals at the specified frequency and clock cycles between the various Bluetooth devices.
The Bluetooth core protocols further provide two different types of physical links with corresponding baseband packets. A synchronous connection-oriented (SCO) physical link and an asynchronous connectionless (ACL) physical link may be implemented in a multiplexed manner on the same RF link. ACL packets are used for data only while the SCO packets may contain audio, as well as a combination of audio and data. All audio and data packets can be provided with different levels of error correction and may also be encrypted if required. Special data types, including those for link management and control messages, are transmitted on a specified channel.
There are other protocols and types of networks being implemented and that may be used with the network of FIG. 1A. For example, wireless networks that comport with service premises-based Wireless Local Area Network (WLAN) communications, e.g., IEEE 802.11a and IEEE 802.11b communications, and ad-hoc peer-to-peer communications, e.g., Bluetooth (as described above). In a WLAN system, the structure would be similar to that shown in FIG. 1A, but, instead of base stations 102 through 112, the WLAN system would include a plurality of Wireless Access Points (WAPs). Each of these WAPs would service a corresponding area within the serviced premises and would wirelessly communicate with serviced wireless devices. For peer-to-peer communications, such as those serviced in Bluetooth applications, the RF transceiver of the present invention would support communications between peer devices, e.g., mobile computer 124 and wireless handset device 126. The fast growth of the mobile communications market and for networks as shown in FIG. 1A require the development of multi-band RF transceivers that are small in size, low in cost, and have low power consumption. These RF transceivers should be suitable for a high level of system integration on a single chip for reduced cost and miniaturized mobile device size. Low power consumption is very critical for increasing mobile device battery life, especially for mobile devices that include small batteries.
Generally, Bluetooth facilitates the fabrication of a low-cost and low-power radio chip that includes some of these protocols described herein. The Bluetooth protocol operates in the unlicensed 2.4 GHz Industrial Scientific Medical (ISM) band and, more specifically, transmits and receives on 79 different hop frequencies at a frequency in the approximate range of 2400 to 2480 MHz, switching between one hop frequency to another in a pseudo-random sequence. Bluetooth, in particular, uses Gaussian Phase Shift Keyed (GFSK) modulation. Its maximum data rate is approximately 721 kbits/s and the maximum range is up to 20-30 meters.
Even though Bluetooth has a much lower range and throughput than other known systems, its consequently significantly reduced power consumption means it has the ability to be much more ubiquitous. It can be placed in printers, keyboards, and other peripheral devices, to replace short-range cables. It can also be placed in pagers, mobile phones, and temperature sensors to allow information download, monitoring and other devices equipped with a Bluetooth access point. Nonetheless, it is advantageous to improve the low power consumption of Bluetooth devices to improve battery life for portable applications.
Similarly, wireless LAN technologies (such as those formed to be compatible with IEEE 802.11b) are being designed to complement and/or replace the existing fixed-connection LANs. One reason for this is that the fixed connection LANs cannot always be implemented easily. For example, installing wire in historic buildings and old buildings with asbestos components makes the installation of LANs difficult. Moreover, the increasing mobility of the worker makes it difficult to implement hardwired systems. In response to these problems, the IEEE 802 Executive Committee established the 802.11 Working Group to create WLAN standards. The standards specify an operating frequency in the 2.4 GHz ISM band.
The first IEEE 802.11 WLAN standards provide for data rates of 1 and 2 Mbps. Subsequent standards have been designed to work with the existing 802.11 MAC layer (Medium Access Control), but at higher frequencies. IEEE 802.11a provides for a 5.2 GHz radio frequency while IEEE 802.11b provides for a 2.4 GHz radio frequency band (the same as Bluetooth). More specifically, the 802.11b protocol operates in the unlicensed 2.4 GHz ISM band. Data is transmitted on Binary Phase Shift Keyed (BPSK) and Quadrature Phase Shift Keyed (QPSK) constellations at 11 Mbps. 802.11b data rates include 11 Mbits/s, 5.5, 2 and 1 Mbits/s, depending on distance, noise and other factors. The range can be up to 100 m, depending on environmental conditions.
Because of the high throughput capability of 802.11b devices, a number of applications are more likely to be developed using 802.11b for networks such as that shown in FIG. 1A. These technologies will allow the user to connect to wired LANs in airports, shops, hotels, homes, and businesses in networks even though the user is not located at home or work. Once connected the user can access the Internet, send and receive email and, more generally, enjoy access to the same applications the user would attempt on a wired LAN. This shows the success in using wireless LANs to augment or even replace wired LANs.
The RF circuitry of the present invention is designed to satisfy at least some of the above mentioned standard-based protocols and may be formed in any of the subscriber units 114 through 128, base stations 102 through 112 or in any other wireless device, whether operating in a cellular system or not. The RF circuitry of the present invention includes low power designs that utilize CMOS technology and that support the defined protocols in a more efficient manner. Thus, for example, the teachings of the present invention may be applied to wireless local area networks, two-way radios, satellite communication devices, or other devices that support wireless communications. One challenge, however, with CMOS design in integrated circuits is that they typically utilize voltage sources having low values (e.g., 3 volts) and are generally noisy. It is a challenge, therefore, to develop receiver and transmitter circuitry that have full functionality while meeting these lower power constraints and while providing good signal quality. The system of FIGS. 1A and 1B include the inventive gain circuitry gain control disclosed herein to provide discrete gain steps to avoid over amplification and under amplification of a down converted signal prior to delivery to a baseband processor.
FIG. 1B is a block diagram generally illustrating the structure of a wireless device 150 constructed according to the present invention. The general structure of wireless device 150 will be present in any of wireless devices 114 through 128 illustrated in FIG. 1A. Wireless device 150 includes a plurality of host device components 152 that service all requirements of wireless device 150 except for the RF requirements of wireless device 150. Of course, operations relating to the RF communications of wireless device 150 will be partially performed by host device components 152.
Coupled to host device components 152 is a Radio Frequency (RF) interface 154. RF interface 154 services the RF communications of wireless device 150 and includes an RF transmitter 156 and an RF receiver 158. RF transmitter 156 and RF receiver 158 both couple to an antenna 160. One particular structure of a wireless device is described with reference to FIG. 2. Further, the teachings of the present invention are embodied within RF transmitter 156 of RF interface 154.
The RF interface 154 may be constructed as a single integrated circuit. However, presently, the RF interface 158 includes an RF front end and a baseband processor. In the future, however, it is anticipated that many highly integrated circuits, e.g., processors, system on a chip, etc., will include an RF interface, such as the RF interface 154 illustrated in FIG. 1B. In such case, the receiver structure of the present invention described herein may be implemented in such devices.
FIG. 2 is a block diagram illustrating a subscriber unit 202 constructed according to the present invention. Subscriber unit 202 operates within a cellular system, such as the cellular system described with reference to FIG. 1A. Subscriber unit 202 includes an RF unit 204, a processor 206 that performs baseband processing and other processing operations, and a memory 208. RF unit 204 couples to an antenna 205 that may be located internal or external to the case of subscriber unit 202. Processor 206 may be an Application Specific Integrated Circuit (ASIC) or another type of processor that is capable of operating subscriber unit 202 according to the present invention. Memory 208 includes both static and dynamic components, e.g., Dynamic Random Access Memory (DRAM), Static Random Access Memory (SRAM), Read Only Memory (ROM), Electronically Erasable Programmable Read Only Memory (EEPROM), etc. In some embodiments, memory 208 may be partially or fully contained upon an ASIC that also includes processor 206. A user interface 210 includes a display, a keyboard, a speaker, a microphone, and a data interface, and may include other user interface components, as well. RF unit 204, processor 206, memory 208, and user interface 210 couple via one or more communication buses or links. A battery 212 is coupled to, and powers, RF unit 204, processor 206, memory 208, and user interface 210.
RF unit 204 includes the RF transceiver components and operates according to the present invention to adjust the amplification or gain level of a programmable gain amplifier prior to delivery of a down converted signal to a baseband processor. More specifically, RF unit 204 includes the gain control circuitry as described herein receives detected peak values from an analog peak detector and generates gain control commands responsive thereto. Further, the RF unit 204 provides circuitry that generates a received signal strength indication as a part of providing gain control commands thereby eliminating a need for a discrete received signal strength indicator circuit for determining the signal strength of a received and down converted RF signal. The structure of subscriber unit 202, as illustrated, is only one particular example of a subscriber unit structure. Many other varied subscriber unit structures could be operated according to the teachings of the present invention. Further, the principles of the present invention may be applied to base stations, as are generally described with reference to FIG. 1A.
FIG. 3 is a functional schematic diagram of an amplifier system. Traditionally, the gain of an operational amplifier is determined by the value of the ratio of a pair of resistors. Specifically, an input resistor is coupled to a negative input of an operational amplifier (Op Amp) and another resistor is placed across the input and output of the Op Amp in a feedback loop. The ratio of the resistive values then determines the gain. Here in the example of FIG. 3, a resistor 304 is coupled the input of an Op Amp 308. A resistor 312 is coupled across the output of Op Amp 308 and to the input node of Op Amp 308 that is also coupled to resistor 304. The gain of the amplifier of FIG. 3, meaning the ratio of the output signal to the input signal, is defined by the ratio of resistor 312 to resistor 304. Typically, to make the gain of the amplifier of FIG. 3 programmable, a variable resistance is coupled to the input of Op Amp 308 so that the ratio of resistor 312 to resistor 304 may be modified.
FIG. 4 is an amplifier formed to have programmable gain. Referring now to FIG. 4, it may be seen that Op Amp 308 has resistor 312 coupled from its output to its negative input. Resistor 304, however, is replaced by a resistor 404 that is in parallel with the combination of a resistor 408 and a switch 412. As is understood, the branch including resistor 408 and switch 412 may represent a plurality of such resistor and switch combinations so that the overall resistance on the input side of Op Amp 308 may be modified, thereby modifying the ratio of resistor 312 to the total resistance represented by resistors 404 and 408 of FIG. 4.
The example of FIG. 4 is one that may be implemented with any number of resistors on the input side coupled to a switch, wherein the number of resistors is a function of the number of gain settings that a designer wishes to implement in his or her system. The solution of FIG. 4 has two inherent problems, however. First, resistors formed within integrated circuit devices require a relatively large amount of semiconductor area and, accordingly, are not design efficient. Second, the resistance of the switches is not constant and can vary as much as 20% due to the switches being typically formed by MOSFET devices. Accordingly, because the resistance in any one branch is a function of the MOSFET switch, as well as the resistor which itself may have a 20% fluctuation in resistive value, the gain accuracy can be significantly degraded to a point that the amplifier does not meet design specifications. Because many radio transceiver devices have multiple amplification stages, however, it is important to control the amount of amplification at various stages of the radio transceiver so as to avoid saturation and maintain satisfactory operation. Thus, it is desirable to accurately predict output gain.
FIG. 5 is a functional schematic block diagram of an RF processing unit of a radio transceiver formed according to one embodiment of the present invention. An RF receiver unit initially receives a radio frequency signal at a receiver/low noise amplifier (LNA) 504 that is coupled to receive wireless communications by way of an antenna. As is known by those of average skill in the art, radio communications typically employ one of many different modulation techniques, including Quadrature Phase Shift Keying (QPSK). In the described embodiment, QPSK modulation is utilized. Accordingly, receiver/LNA 504 produces RF with I and Q components for processing. Within each of the branches carrying the I and Q components, the signal is produced to a mixer oscillator that down converts the received signal from radio frequencies to a baseband channel and extracts the I and Q components.
In the described embodiment of the invention, the radio transceiver is formed to satisfy 802.11b design requirements. Accordingly, the input signal is tuned to a 2.4 GHz frequency of oscillation and then is down converted to baseband. Mixers 508A and 508B produce the down converted signal to a low pass filter 512A and a low pass filter 512B for the I and Q modulated channels, respectively.
As is known, low pass filters 512A and 512B are for blocking all communication signals above a specified frequency. The outputs of low pass filters 512A and 512B are then produced to programmable gain amplifiers (PGAs) 516A and 516B. The outputs of PGAs 516A and 516B are adjusted by gain control circuitry 520 to produce approximately 0 dBm of swing for the amplified signal that is then transmitted into baseband processing circuitry. In the described embodiment, the analog-to-digital converters (ADCs) are within the baseband processing circuitry external to the IF radio integrated circuit.
Each of the components in this path thus far, namely, receiver/LNA 504, mixers 508A and 508B, and low pass filters 512A and 512B, add gain to the received signals. Because the power of the received signal can vary dramatically, however, the gain of the programmable gain amplifier is adjusted so that the gain of the output signal being provided to the baseband radio circuitry is of a constant level. Unfortunately, many common programmable gain amplifiers provide a very limited number of amplification steps, if any. Accordingly, the present invention includes a programmable gain amplifier whose gain can be incremented in 3 dB increments from 0 dB to 45 dB. Additionally, the invention includes circuitry to determine an appropriate amount of gain that should be added to the signal received by the PGA 516A so that it is amplified up to 0 dBm with a tolerance of +/−2 dBm.
FIG. 6 is a functional schematic diagram illustrating one embodiment of the programmable gain amplifiers used in the present invention. More specifically, as may be seen, a PGA, for example, PGA 516A of FIG. 5, includes four amplification stages that collectively provide up to 45 dB of gain. An amplifier 604 provides either 0 or 12 dB of gain. Its output is produced to an amplifier 608 that also produces 0 or 12 dB of gain. The output of amplifier 608 is then provided to an amplifier 612 that, again, provides either 0 or 12 dB of gain. The output of amplifier 612 is then provided an amplifier 616 that provides gain in finer increments, namely, 3 dB. Thus, amplifier 616 provides either 0 dB, 3 dB, 6 dB or 9 dB of gain.
Accordingly, by selecting which of the four amplifiers 604, 608, 612 or 616 is providing gain, and in the case of amplifier 616 how much gain is generated therefrom, the amount of gain provided collectively can be varied in incremental steps of 3 dB all the way from 0 dB of gain to 45 dB of gain (16 logic states). In general, each of the amplifiers 604, 608 and 612 receive a binary signal specifying whether it should provide 0 dB of gain or 12 dB of gain. Amplifier 616, however, receives a four-state signal in one embodiment of the invention, which specifies whether it is to provide 0 dB, 3 dB, 6 dB or 9 dB of gain. As is known by those of average skill in the art, four bits is sufficient to generate the required 16 logic states for this embodiment of the invention. A plurality of logic gates may be used to generate the required amplifier control signals from the four bits as generally described.
FIG. 7 is a functional schematic diagram of an amplifier that provides gain in 3 dB steps. More specifically, the amplifier of FIG. 7 may be used for providing the functionality described in association with amplifier 616 of FIG. 6. As is known by those of average skill in the art, the gain of an operational amplifier is the ratio of the resistance in the feedback loop divided by the resistance at the input stage. Accordingly, the amount of resistance provided by a second group of MOSFETs 712 that are coupled to the input of an Op Amp 708 is divided into the resistance provided by a first plurality of MOSFETs 704 that are coupled in the feedback loop of Op Amp 708.
In the described embodiment, the first group of MOSFETs 704 comprise five matched MOSFETs that are coupled in parallel within the feedback loop of Op Amp 708 and are shown as one MOSFET 716 coupled to operate as a resistor when biased in its linear region.
In the second group of MOSFETs 712, a total of fourteen MOSFETs are coupled in parallel to provide the incremental gain steps for the amplifier of FIG. 7. A first group of five transistors, which is represented by MOSFET 720, is biased to ground and each of the five transistors is always biased in the linear region. The remaining MOSFETs of the second group of MOSFETs 712, however, are selectively biased so as to modify the gain of the amplifier of FIG. 7.
More specifically, two transistors, shown as MOSFET 724, are biased by a first selectable bias voltage S1. Similarly, three transistors, shown as MOSFET 728, are all biased by a second selectable bias voltage S2. Finally, four transistors, represented as MOSFET 732, are biased by a third selectable bias voltage S3. Thus, the five transistors represented by MOSFET 720, will always be on, while the MOSFETs that receive bias voltages S1, S2 and S3, shown herein as MOSFET 724, 728 and 732, respectively, may be selectively biased according to the desired output gain level.
As is understood by those of average skill in the art, the gain of the amplifier (in dB) in FIG. 7 is defined by the equation of 20 log (R2/R1). In this example, R1 comprises the transistors of the second group of MOSFETs 712, and R2 comprises the first group of MOSFETs 704. Because the gain is a function of the logarithm of the resistive values R2l /R1, the gain is modified for the amplifier of FIG. 7 to create the 3 dB steps by modifying the resistive ratios according to the biasing of the transistors in the second group of MOSFETs 712.
For example, if MOSFETs 724, 728 and 732 are biased to an off state, then the gain is 0 dB because the log of the resistance of the five MOSFETs 716 divided by the resistance of five matched MOSFETs 720 is equal to 0 because the log of 1 is equal to 0. To provide 3 dB of gain, seven of the transistors from the second group of MOSFETs 712 must be biased in the linear region. This is because 20 log (7/5) is equal to 2.92 dB, which is approximately equal to 3 dB. Similarly, for 6 dB of gain, ten of the MOSFETs 712 must be biased in the linear region because 20 log (10/5) is equal to 6.02 dB, which is approximately equal to 6 dB. Finally, for 9 dB of gain, all fourteen MOSFETs of the MOSFETs 712 must be biased in the linear region because 20 log (14/5) is equal to 8.94 dB, which is approximately equal to 9 dB.
Thus, to achieve the desired gain steps, the fourteen MOSFETs of the second group of MOSFETs 712, must be operationally biased so that either 5, 7, 10 or 14 transistors are on according to the desired step size. Moreover, as was described before, the five MOSFETs represented by MOSFET 720 are always biased in the linear region by virtue of the gate terminal coupled to ground. Thus, for 0 dB of gain, the remaining transistors are merely biased off by the logic signals. Whenever 3, 6 or 9 dB of output is desired, logic signal S1 is used to operationally bias the MOSFETs represented by MOSFETs 724. Whenever 6 dB of gain is desired, both S1 and S2 are used to bias the transistors represented by MOSFETs 724 and 728 in the linear region. Finally, if all fourteen transistors were to be biased on to provide 9 dB of gain, then S1, S2 and S3 are all set to bias the MOSFETs represented by MOSFETs 724, 728 and 732 of the MOSFETs 712 of FIG. 7.
FIGS. 6 and 7 illustrate the operation of the programmable gain amplifier itself according to one embodiment of the invention. Additionally, however, one aspect of the present embodiment of the invention is the control of the programmable gain amplifier to achieve the desired 3 dB increment steps.
FIG. 8 is a functional block diagram that illustrates one embodiment of the present invention for controlling the programmable gain amplifiers for the I and Q modulated channels. With respect to the I channel, PGA 516A of FIG. 5 receives the I channel communication signals for amplification. The output of PGA 516A is then provided to an amplitude level detector 804. Similarly, PGA 516B receives the communication signals over the Q channel for amplification and produces an amplified output to an amplitude level detector 804. Amplitude level detector 804 provides an output that may be represented by the square root of the sum of the I channel amplitude squared and the Q channel amplitude squared. This output signal represents a peak level of the I and Q modulated channels. Amplitude level detector 804 is coupled to receive the output of PGA 516B, as well as PGA 516A, to produce the amplitude peak detection. Based upon the detected gain level of the I and Q modulated channels, amplitude level detector 804 produces an indication to a gain control logic block 808 which responds by generating control commands to PGA 516A and PGA 516B to increment or decrement the output gain level. In the described embodiment of the invention, gain control logic block 808 includes a 4-bit counter 812 that is driven by an external clock signal for producing the increment and decrement control bits to PGAs 516A and 516B.
FIG. 9 is a functional schematic block diagram that illustrates an analog peak detector circuit formed according to one embodiment of the present invention. PGAs 904 and 908 each produce an amplified output signal. The output of PGA 904 is a differential output signal that may be represented by (a)sin (ωt). The output of PGA 908 is a differential output signal that may be represented by (a)cos (ωt). The outputs of PGAs 904 and 908 are fed into a peak detector 912 formed according to the described embodiments of the invention. Responsive to receiving the outputs of PGAs 904 and 908, peak detector 912 determines the peak amplitude of the outputs of PGAs 904 and 908, provides the determined output to a control circuit 916 which then generates control signals back to PGAs 904 and 908 to reduce or increase the output gain therefrom.
In the described embodiment of the present invention, peak detector 912 comprises analog circuitry for fast determination of the peak amplitude of the outputs of the PGAs 904 and 908. In general, peak detector 912 takes advantage of a square law current to voltage function for MOS devices to determine that peak of the amplified signal. By utilizing a circuit that takes advantage of the square law function for MOS devices, the present invention is able to square the received inputs, thereby eliminating the sine and cosine elements of the signals because of the known equation that holds that sin2+cos2=1. Thus, by also taking the square root of the sum of the squared signals, the peak amplitude “a” of the received quadrature signals may be determined.
FIG. 10 is a functional schematic block diagram that illustrates one embodiment of the present invention. The PGAs 516A and 516B described above, may be seen in the schematic block diagrams of FIG. 8 and FIG. 9. A peak amplitude detector 1004 is coupled to provide an indication of a detected peak amplitude to gain control circuitry 1006. Gain control circuitry 1006 includes a pair of comparators 1008 and 1012. Peak amplitude detector 1004 is coupled to receive the I and Q channel amplified outputs of PGAs 516A and 516B, respectively. Peak amplitude detector 1004 provides an output that represents the square root of the sum of the squared amplitude components of each of the I and Q channels (an overall peak value). The peak value from peak amplitude detector 1004 is provided to comparators 1008 and 1012. Comparator 1008 compares the output of peak amplitude detector 1004 to a low reference voltage value while comparator 1012 compares the output of peak amplitude detector 1004 to a high reference voltage value.
With respect to comparator 1008, if the peak amplitude received from peak amplitude detector 1004 is below the low reference voltage, it generates a signal to 4-bit counter 1016 to increment the 4-bit count value and, thus, increase the gain of PGAs 516A and 516B. Similarly, comparator 1012 will generate a signal to decrease the gain if the detected peak amplitude from peak amplitude detector 1004 is higher than the high reference voltage.
Correspondingly, the gain control circuitry 1006, and, more specifically, the 4-bit counter 1016 generates control signals to PGAs 516A and 516B to increase or decrease the gain. As was described earlier, each of the PGAs 516A and 516B will amplify a signal up to 45 dB in the described embodiment of the invention. 4-bit counter 1016 will set the four control bits to adjust the gain, as has been described herein, in 3 dB increments in the described embodiment of the invention.
The inventor herein has further recognized that the output of 4-bit counter 1016 also describes the inverse of the signal strength for the received signal. Accordingly, by adding an inverter 1020 to the output of 4-bit counter 1016, the output may serve as a received signal strength indicator (RSSI) that is common in these types of radio circuits.
Generally, an RSSI is coupled to the input stage of the programmable gain amplifiers, such as PGAs 516A and 516B, to provide an indication of the received signal strength. The circuitry used to control the gain of PGAs 516A and 516B may also be used to provide an indication of the received signal strength in place of a received signal strength indicator by merely inverting the output of 4-bit counter 812. Accordingly, the output of 4-bit counter 1016 is provided to inverter 1020, whose output serves as the output of a received signal strength indicator.
As is understood by those of average skill in the art, typical radio transceiver designs include two RSSIs so that a determination may be made about the amount of interference as well as the signal strength of the received signal. This information is used for controlling gain at the various stages of the radio transceiver. RSSI 1 (a first RSSI) is usually coupled at the input of a low pass filter to measure the signal strength of the combination of the received signal and interferers in the same and in adjacent channels. RSSI 2 (the second RSSI) is usually coupled at the output of the low pass filters and at the input of the PGA and is used to measure the signal strength of the desired signal. The output of inverter 1016 serves to provide the received signal strength indicator of RSSI 2, meaning the signal strength of the signal only.
The invention disclosed herein is susceptible to various modifications and alternative forms. Specific embodiments therefore have been shown by way of example in the drawings and detailed description. It should be understood, however, that the drawings and detailed description thereto are not intended to limit the invention to the particular form disclosed, but on the contrary, the invention is to cover all modifications, equivalents and alternatives falling within the spirit and scope of the present invention as defined by the claims.

## Claims

1. A radio receiver formed in an integrated circuit, comprising:
a receiver and low noise amplifier circuit (Rx/LNA) coupled to receive an RF signal;
a mixer coupled to receive an amplified RF signal of the Rx/LNA, the mixer for down converting the received amplified RF signal to produce a down converted signal having quadrature and in-phase components;
a programmable gain amplifier (PGA) block for providing coarse and fine incremental gain steps based upon a gain control signal of the down converted signal, wherein the PGA block further includes:
an operational amplifier (Op Amp);
a first group of MOSFETs coupled in parallel in a feedback loop of the Op Amp between an output and negative input of the Op Amp; and
a second group of MOSFETs coupled to the negative input of the Op Amp wherein the MOSFETs of the first and second groups are matched and further wherein a number of selectable transistors in the second group of MOSFETs is chosen so that increments of gain in terms of dB may be achieved from zero dB to a dB value that is equal to or less than a gain step size produced by the PGA block for providing a coarse incremental step; and

gain control logic for generating the gain control signal based upon peak values of quadrature and in-phase down converted signals.

2. The radio receiver of claim 1 operable to determine a square root of a sum of squares of the quadrature and in-phase down converted signals as a part of generating the gain control signal.

3. The radio receiver of claim 1 wherein at least one of the MOSFETs of the first group of MOSFETs is permanently coupled to operate in a linear mode whenever the transceiver is in an operational state.

4. The radio receiver of claim 1 wherein the second group of MOSFETs comprises at least five MOSFETs that are permanently coupled to operate in a linear mode of operation whenever the transceiver is in an operational state.

5. The radio receiver of claim 4 wherein the second group of MOSFETs comprises at least fourteen MOSFETs.

6. The radio receiver of claim 4 wherein a plurality of MOSFETs are selectable to create a gain step size of a specified amount in terms of dB.

7. The radio receiver of claim 6 wherein the specified amount is 3 dB.

8. The radio receiver of claim 7 wherein the step sizes include approximately 0 dB, 3 dB, 6 dB and 9 dB.

9. A radio receiver formed within an integrated circuit, comprising:
a low noise amplifier coupled to receive a wireless communication signal;
a mixer for tuning and down converting the received wireless communication signal produced from the low noise amplifier;
a low pass filter coupled to receive a down converted signal produced from the mixer;
a programmable gain amplifier operably coupled to amplify an output produced by the low pass filter, further comprising:
a plurality of selectable coarse gain amplifiers coupled in series for providing gain in coarse increments; and
at least one programmable fine gain amplifier coupled in series with the selectable coarse gain amplifiers;

an amplitude level detector for detecting the amplitude of an amplified signal produced by the plurality of selectable coarse gain amplifiers and the at least one programmable fine gain amplifier, the amplitude level detector operable to calculate a square root of squared in-phase and quadrature components of down-converted signals produced by the mixer as a part of detecting the amplitude;
logic circuitry coupled to receive a detected amplitude from the amplitude level detector, the logic circuitry for providing control signals for incrementing or decrementing the amplitude of the amplified signal; and
a counter coupled to receive the control signals and to the programmable gain amplifier to produce gain increase and decrease signals to the programmable gain amplifier.

10. The radio receiver of claim 9 wherein the logic circuitry further includes a first comparator coupled to receive the detected amplitude and a low reference voltage.

11. The radio receiver of claim 10 wherein the first comparator generates a control command to the counter to increase the voltage amplitude if the detected peak amplitude is lower than the low reference voltage.

12. The radio receiver of claim 11 wherein the logic circuitry further includes a second comparator coupled to receive the detected amplitude and a high reference voltage.

13. The radio receiver of claim 12 wherein the second comparator generates a control command to the counter to decrease the voltage amplitude if the detected peak amplitude is higher than the high reference voltage.

14. A method for producing an indication of a received signal strength, comprising:
receiving, at a peak amplitude detector, a signal having an amplitude;
producing an indication of a peak amplitude of the signal wherein the indication is a square root of a sum of squares of in-phase and quadrature phase components of the signal having an amplitude;
determining whether the amplitude is too high if the amplitude is greater than a first threshold value;
determining whether the amplitude is too low if the amplitude is less than a second threshold value;
incrementing and decrementing a counter according to whether the amplitude is too high or too low; and
receiving and inverting a counter output received from the counter to produce a received signal strength indication.

15. The method of claim 14 wherein the received signal is received from an output of a low pass filter.

16. The method of claim 14 wherein the step of producing an indication of a peak amplitude includes receiving a first signal having an amplitude having a first phase.

17. The method of claim 16 wherein the step of producing an indication of the peak amplitude includes receiving a second signal having an amplitude having a second phase.

18. The method of claim 14 wherein the step of producing an indication of the peak amplitude includes producing an output signal whose value is represented by a square root of the sum of the squares of a magnitude indication of the first and second signals.

19. The method of claim 14 wherein the step of receiving the counter output comprises receiving a 4-bit counter signal and inverting the values of each bit of the four bits.