# Patent

## Title
Power converter employing a tapped inductor and integrated magnetics and method of operating the same

## Abstract

A power converter employing a magnetic device including a magnetic core and first and second tapped windings. Each winding of the magnetic device has first and second winding sections, respectively, coupled to the magnetic core, wherein the first and second winding sections are formed from a formed and folded planar conductive sheet. The magnetic core includes a common leg, a plurality of outer legs, and a top plate. The first and second winding sections are constructed with a formed and folded planar conductive sheet formed with structures usable as pins in a through-hole, surface-mount, or combined printed wiring board assembly process. The outer legs at either end of the magnetic core are positioned back from magnetic core ends, and the magnetic core is formed with a stepped structure to accommodate winding pins in a surface-mount printed wiring board assembly process.

## Background
This application is a continuation in part of patent application Ser. No. 11/655,334, entitled “Power Converter Employing a Tapped Inductor and Integrated Magnetics and Method of Operating the Same,” filed Jan. 19, 2007, now U.S. Pat. No. 7,298,118, which is a continuation of patent application Ser. No. 11/361,742, entitled “Power Converter Employing a Tapped Inductor and Integrated Magnetics and Method of Operating the Same,” filed on Feb. 23, 2006, now U.S. Pat. No. 7,176,662, which claims the benefit of U.S. Provisional Application No. 60/655,707, entitled “Integrated Multi-Phase Tapped Inductor Buck Converter, Integrated Magnetic Device, and Method of Operation,” filed on Feb. 23, 2005, and claims the benefit of provisional application Ser. No. 60/905,527, entitled “Power Converter Employing a Tapped Inductor and Integrated Magnetics and Method of Operating the Same,” filed on Mar. 7, 2007, all of which are incorporated herein by reference.


TECHNICAL FIELD
The present invention is directed, in general, to power electronics and, in particular, to power converters using a tapped inductor and integrated magnetics, and methods of operating the same.
BACKGROUND
A switched-mode power converter (also referred to as a “power converter”) is an electronic power processing circuit that converts an input voltage waveform into an output voltage waveform. The waveforms are typically, but not necessarily, dc waveforms, controlled by periodically switching power switches or switches coupled to an inductive circuit element. The switches are generally controlled with a conduction period “D” referred to as a “duty cycle.” The duty cycle is a ratio represented by the conduction period of a switch to a switching period thereof. Thus, if a switch conducts for half of the switching period, the duty cycle for the power switch would be 0.5 (or 50 percent).
Feedback controllers associated with power converters manage an operation thereof by controlling the conduction period of a switch employed therein. Generally, a feedback controller is coupled to an output of a power converter in a feedback loop configuration (also referred to as a “control loop” or “closed control loop”) to regulate an output characteristic of the power converter such as an output voltage. A switched-mode power converter typically receives a dc input voltage Vin from a source of electrical power at input nodes thereof and provides a regulated output voltage Vout at output nodes thereof to power, for instance, a microprocessor coupled to the output nodes of the power converter.
Switched-mode power converters are key components in many commercial and military systems for the conversion, control and conditioning of electrical power, and often govern performance and size of the end system. Power density, efficiency and reliability are key metrics used to evaluate power converters. Magnetic devices including transformers and inductors used within power converters contribute a significant percentage to the volume and weight and, hence, determine power density, efficiency, and reliability.
An integrated magnetic device (also referred to as “integrated magnetics”) provides a technique to combine multiple inductors and/or transformers in a single magnetic core and is specifically amenable to interleaved power converter topologies where the input or output current is shared among multiple inductors. Integrated magnetics offers several advantages such as improved power density and reduced cost due to the elimination of separate magnetic components, reduced switching ripple in inductor currents, and higher efficiency due to reduced magnetic core and copper losses.
For applications where higher currents (typically greater than 50 amps (“A”)) are required at low (typically less than 3.3 volts (“V”)) to moderate (typically about 12 V) voltages at high efficiency and power density, a two-phase interleaved power converter might be inadequate to meet switching ripple or response time specifications on inductor currents and output voltage. A larger output capacitor can reduce the output ripple voltage, but will increase the volume and weight of the power converter and result in sluggish transient response to dynamic load conditions. Multi-phase, interleaved power converters beyond the present two-phase designs may advantageously be employed for such applications. Utilizing multiple discrete magnetic cores (e.g., E-cores) to implement multi-phase interleaved power converters and simply paralleling multiple power converters, however, increases component count and interconnect losses, resulting in poor power density and efficiency.
To meet response time requirements in systems operating with high current and very low bias voltage, such as 1.5 volts or lower, it is often necessary to place a voltage regulator module in the form of a dedicated dc-dc converter in close proximity to the load. In this manner, an accurate supply voltage can be delivered to a sensitive load such as a microprocessor. Many voltage regulator modules in use today are based on a multi-phase buck power converter. In a multi-phase buck power converter, the duty cycle D equals the ratio of the output voltage Vout to the input voltage Vin thereof. Microprocessors for desktop computers, workstations, and low-end servers often employ voltage regulator modules to work with a 12 volt input. In laptop computers, the voltage regulator modules often directly convert the battery charger voltage of 16 to 24 volts down to the microprocessor voltage of 1.5 volts. For future microprocessors, the supply voltage is expected to decrease to below one volt to further reduce power dissipation and to accommodate the fine line geometries used to form the integrated circuits that form microprocessors and the like.
For the aforementioned applications, a multi-phase power converter (e.g., a multi-phase buck power converter) is often employed to operate at very small duty cycles to regulate a low output voltage from a substantially higher input voltage. At very small duty cycles, both the transient response and the efficiency of a multi-phase power converter may be compromised. To improve power conversion efficiency without compromising transient response, alternative topologies that extend duty cycles to a higher level in such applications with a high ratio of input to output voltage would be advantageous.
An additional limitation to using magnetic cores (e.g., E-cores) for high current applications is the detrimental effects of fringing magnetic flux due to the limited cross-sectional area of a gapped center leg of the magnetic device. Fringing magnetic flux represents the magnetic flux component that strays away from the main magnetic path and spills into a core window, inducing eddy currents in the windings of the magnetic device. This results in increased losses (e.g., denoted by I2R, wherein “I” represents the current and “R” represents the resistance) in the windings and reduced efficiency. To reduce the induction of eddy currents due to fringing magnetic flux, the windings are placed a safe distance from an air gap, resulting in poor conductor utilization of the core window area. In addition, fringing magnetic flux represents a loss of inductance which results in increased switching ripple in the winding currents, leading to higher losses and poorer efficiencies.
Multi-phase, interleaved power converter topologies can thus provide highly desirable power conversion designs, not only for their small size, but also for their ability to provide fast response times for a controller regulating the output voltage thereof with minimal output ripple voltage. A power converter that combines the advantages of an interleaving, multi-phase power converter topology with a circuit arrangement that can be implemented to operate at higher levels of duty cycle while substantially reducing the magnetic flux variation in a magnetic core, and with integrated magnetics is not presently available for the more severe applications that lie ahead.
Accordingly, what is needed in the art is a power converter topology that employs switches that can operate with higher levels of duty cycle that can reduce magnetic flux changes in a magnetic core of a magnetic device thereof, and overcomes the deficiencies in the presently available power converters.
SUMMARY OF THE INVENTION
These and other problems are generally solved or circumvented, and technical advantages are generally achieved, by advantageous embodiments of the present invention that include a power converter employing a magnetic device and method of operating the same. In one embodiment, the magnetic device includes a magnetic core and first and second tapped windings, each winding having first and second winding sections, respectively, coupled to the magnetic core. In an advantageous embodiment, the first and second winding sections are formed from a formed and folded planar conductive sheet. In an advantageous embodiment, the first and second winding sections are constructed with a formed and folded planar conductive sheet formed with structures usable as pins in a through-hole printed wiring board assembly process. In a further advantageous embodiment, the first and second winding sections are constructed with a formed and folded planar conductive sheet formed with structures usable as pins in a surface-mount printed wiring board assembly process.
The foregoing has outlined rather broadly the features and technical advantages of the present invention in order that the detailed description of the invention that follows may be better understood. Additional features and advantages of the invention will be described hereinafter which form the subject of the claims of the invention. It should be appreciated by those skilled in the art that the conception and specific embodiment disclosed may be readily utilized as a basis for modifying or designing other structures or processes for carrying out the same purposes of the present invention. It should also be realized by those skilled in the art that such equivalent constructions do not depart from the spirit and scope of the invention as set forth in the appended claims.



BRIEF DESCRIPTION OF THE DRAWINGS
For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which:
FIG. 1 illustrates a schematic diagram of an embodiment of a power converter constructed according to the principles of the present invention;
FIG. 2 illustrates a side view of an embodiment of a magnetic device constructed according to the principles of the present invention;
FIG. 3 illustrates a reluctance model of a magnetic device constructed according to the principles of the present invention;
FIGS. 4 and 5 illustrate schematic diagrams demonstrating exemplary operations of an embodiment of a power converter constructed according to the principles of the present invention;
FIG. 6 illustrates a graphical representation of the change of magnetic flux in one leg of a magnetic device in a power converter constructed according to the principles of the present invention;
FIG. 7 illustrates a schematic diagram of another embodiment of a power converter constructed according to the principles of the present invention;
FIG. 8 illustrates a graphical representation of contour plots of the ratio of output voltage to input voltage as a function of duty cycle and the turns ratio of coupled sections of windings for a power converter constructed according to the principles of the present invention;
FIG. 9 illustrates a graphical representation of waveforms for magnetic flux in one leg of a magnetic device and output current from one active phase leg for a power converter constructed according to the principles of the present invention;
FIG. 10 illustrates a schematic diagram of an embodiment of a power converter constructed according to the principles of the present invention;
FIG. 11 illustrates a side view of another embodiment of a magnetic device constructed according to the principles of the present invention;
FIG. 12 illustrates a perspective view of another embodiment of a magnetic device constructed according to the principles of the present invention;
FIG. 13 illustrates a reluctance model of a power converter constructed according to the principles of the present invention;
FIG. 14 illustrates a graphical representation demonstrating ripple cancellation effects for a power converter constructed according to the principles of the present invention; and
FIGS. 15 to 28 illustrate views of further embodiments of magnetic devices constructed according to the principles of the present invention.



DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS
The making and using of the presently preferred embodiments are discussed in detail below. It should be appreciated, however, that the present invention provides many applicable inventive concepts that can be embodied in a wide variety of specific contexts. The specific embodiments discussed are merely illustrative of specific ways to make and use the invention, and do not limit the scope of the invention. Unless otherwise provided, like designators for devices employed in different embodiments illustrated and described herein do not necessarily mean that the similarly designated devices are constructed in the same manner or operate in the same way. It should also be understood that the use of the terms such as “first” and “second” are employed for purposes of explanation and clarity, and reversing such designations when referring to device(s) is well within the broad scope of the present invention.
The invention will be described with respect to an exemplary embodiment in a specific context, namely, a power converter employing a main switch coupled to tapped inductor(s) constructed with, for instance, integrated magnetics. The particular embodiment described herein is a multi-phase, interleaved, dc-dc power converter topology including integrated magnetics to combine multiple tapped inductors in a single magnetic core. The principles of the present invention may be applied to other magnetic device construction arrangements, for example, inclusion of a plurality of magnetic cores in the construction arrangement with multiple tapped inductors, thereby employing less magnetic integration, as well as inclusion of one or more separate inductors not integrated with a tapped inductor.
Referring now to FIG. 1, illustrated is a schematic diagram of an embodiment of a power converter constructed according to the principles of the present invention. The power converter (e.g., a single-phase tapped-inductor buck power converter) includes a main switch Qmain and an auxiliary switch Qaux coupled to single-phase tapped inductor Ltp, which forms an active phase leg thereof. While the main and auxiliary switches Qmain, Qaux are illustrated as metal-oxide semiconductor field-effect transistors (“MOSFETs”), other switches such as insulated-gate bipolar transistors with parallel diodes can be used to advantage. The main and auxiliary switches Qmain, Qaux are assumed to include body diodes that are effectively coupled between the source and drain terminals thereof.
The drain terminal of the main switch Qmain is coupled to a positive terminal of a source of electrical power (represented by a battery) that provides an input voltage Vin for the power converter at input nodes Ni1, Ni2. The source terminal of the main switch Qmain is coupled to a winding of the tapped inductor Ltp. The first and second inductors L1, L2 represent magnetizing inductances of the tapped inductor Ltp. The auxiliary switch Qaux operates as a synchronous rectifier for the power converter. In an alternative embodiment, the auxiliary switch Qaux can be replaced with a diode or with an alternative active rectifying element. A load (represented by a resistor RL) is coupled to output terminals No1, No2 of the power converter. An output capacitor Cout is preferably included in the power converter to filter the output voltage Vout produced at the output terminals No1, No2 of the power converter.
During normal switching operation of the power converter, the main switch Qmain is enabled to conduct during a period designated 0<t<DTS, where TS represents the switching period for the power converter and D represents duty cycle for the main switch Qmain. The auxiliary switch Qaux is enabled to conduct during a complementary period designated DTS<t<TS. Additionally, a brief dead time may be interposed between the conduction periods of the main and auxiliary switches Qmain, Qaux to avoid cross conduction and to accommodate reduced-voltage or zero-voltage switching therefor. Thus, the main and auxiliary switches Qmain, Qaux conduct in a substantially complementary manner. In other embodiments, other control schemes may be used such as providing substantial dead time between the closure of the main and auxiliary switches Qmain, Qaux. If a diode is employed for the auxiliary switch Qaux, the diode typically will be forward biased during the complementary period of the switching cycle, particularly during continuous conduction mode of operation of the power converter.
Turning now to FIG. 2, illustrated is a side view of an embodiment of a magnetic device constructed according to the principles of the present invention. The magnetic device is analogous to the tapped-inductor Ltp illustrated and described with respect to FIG. 1. The tapped inductor is formed using a magnetic core (designated as “CR”), such as a soft ferrite core, including an air gap (designated “GP”) in a magnetic path thereof. A tapped winding Wtp is wound around one leg of the magnetic core CR with primary turns Np in a first section of the tapped winding Wtp and secondary turns NS in a second section of the tapped winding Wtp. A current in either section of the tapped winding Wtp produces a magnetic flux φL in the magnetic core CR and in the air gap GP as indicated therein.
Turning now to FIG. 3, illustrated is a reluctance model of a magnetic device constructed according to the principles of the present invention. More specifically, the illustrated reluctance model represents the tapped inductor Ltp represented in the aforementioned FIGUREs. A magnetomotive force generator (designated “MMFG”) induces magnetomotive force of magnitude [(Np+NS)iL] in a magnetic core and an air gap (see, for instance, FIG. 2). It should be understood that lower case variables used to denote currents and voltages represent instantaneous values while upper case variables denote corresponding average values. A tapped winding (such as the tapped winding Wtp in FIG. 2) is formed with primary and secondary turns represented by (Np+NS), and is assumed to conduct a current “iL” through both of the first and second sections thereof, resulting in the magnetic flux φL represented therein. The reluctance of the magnetic core and the air gap is represented by a resistor with reluctance RC.
Turning now to FIGS. 4 and 5, illustrated are schematic diagrams demonstrating exemplary operations of an embodiment of a power converter constructed according to the principles of the present invention. In particular, the illustrated embodiments demonstrate exemplary operations of the power converter illustrated and described with respect to FIG. 1. In FIG. 4, during a period when the main switch Qmain is enabled to conduct (i. e., during a period represented by 0<t<DTS), a current iL flows from the source of electrical power through the magnetizing inductances represented by first and second inductors L1, L2 of the tapped inductor Ltp toward the output of the power converter. Thus, main switch Qmain is controlled to induce the magnetizing current iL from the source of electrical power into the tapped winding of the active phase leg. The magnetizing current IL is the input current from the source of electrical power when the main switch Qmain is turned on.
In FIG. 5, during the period when the main switch Qmain is not enabled to conduct and the auxiliary switch Qaux is enabled to conduct (i.e., during a complementary period DTS<t<TS), a first current I1 flows through the magnetizing inductance represented by the first inductor L1 and the first section of the tapped winding (with primary turns Np) of the tapped inductor Ltp. To satisfy continuity of current, the first current I1 flowing through the first winding is equal to the magnetizing current iL. A second current I2 flows through the auxiliary switch Qaux through the second section of the tapped winding (with secondary turns NS) of the tapped inductor Ltp. This represents the energy stored in the first inductor L1 due to the current iL. Hence, the second current I2 is equal to the magnetizing current iL scaled by the turns ratio Np/NS. In addition, a third current I3 flows through the magnetizing inductance represented by the second inductor L2. This component is the magnetizing current iL that was flowing through the second inductor L2 when the main switch Qmain was enabled. Finally, an output current Io flows to the load (represented by a resistor RL). Thus, the auxiliary switch Qaux operates to provide an output current Io at an output node No1 via the second section of the tapped winding of the tapped inductor Ltp. The magnitude of the currents I1, I2, I3 can be readily estimated from field continuity equations for a magnetic device, as is well understood in the art. The current iL represents the magnetizing current and is directly proportional to the magnetic flux flowing in the corresponding leg of a magnetic core of the tapped inductor Ltp.
The secondary turns NS of the tapped inductor Ltp typically will have fewer turns than the primary winding Np. Consequently, substantial currents for the second and third currents I2, I3 can be conducted from ground to an output node of the power converter with little power loss. Circuit elements with the same reference designations in the FIGUREs will not be redescribed in the interest of brevity.
Turning now to FIG. 6, illustrated is a graphical representation of the change of magnetic flux in one leg of a magnetic device in a power converter constructed according to the principles of the present invention. More specifically, the graphical representation illustrates the magnetic flux φL flowing in a leg of the magnetic core illustrated and described with respect to FIG. 2 for the exemplary operations illustrated and described with respect to FIGS. 5 and 6. During the portion of a switching cycle when the main switch Qmain is enabled to conduct (i.e., during a period 0<t<DTS), the magnetic flux φL in the magnetic core increases according to the equation




  
    
      
        (
        
          
            N
            p
          
          +
          
            N
            s
          
        
        )
      
      ⁢
      
        
          d
          
            φ
            L
          
        
        
          d
          t
        
      
    
    =
    
      
        V
        
          i
          ⁢
          
            
          
          ⁢
          n
        
      
      -
      
        V
        out
      
    
  
  ,




representing the segment LGA of the magnetic flux waveform, and the expression





  
    
      
        V
        
          i
          ⁢
          
            
          
          ⁢
          n
        
      
      -
      
        V
        out
      
    
    
      
        N
        p
      
      +
      
        N
        s
      
    
  
  ,




representing the slope SLA.

During a complementary period (i.e., DTS<t<TS), the magnetic flux φL in the magnetic core decreases according to the equation




  
    
      
        N
        s
      
      ⁢
      
        
          d
          
            φ
            L
          
        
        
          d
          t
        
      
    
    =
    
      -
      
        V
        out
      
    
  
  ,




representing the segment LGB of the magnetic flux waveform, and the expression





  
    
      -
      
        V
        out
      
    
    
      N
      s
    
  
  ,




representing the slope SLB. During steady-state operation, the magnetic flux φL returns to the same level at the end of a switching cycle as it had at the beginning of the switching cycle.

Turning now to FIG. 7, illustrated is a schematic diagram of another embodiment of a power converter (e.g., a single-phase tapped-inductor buck power converter) constructed according to the principles of the present invention. In the illustrated embodiment, the series position of the main switch Qmain and the first section of the tapped winding (with primary turns Np) of the tapped inductor Ltp are reversed in comparison to their positions indicated in the power converter illustrated and described with respect to FIG. 1. The tapped inductor Ltp is formed from two separated winding sections (i.e., the first and second winding section) and the operation thereof is substantially the same as provided above with respect to the power converter illustrated and described with respect to FIG. 1, et seq.
An equation representing the input-output voltage ratio for a power converter can be readily derived recognizing that the volt-seconds per turn over a switching period during steady-state operation for the tapped inductor Ltp during the period 0<t<DTS, when the main switch Qmain is conducting, and during the complementary period (i.e., DTS<t<TS), when the auxiliary switch Qaux is conducting, are equal and is set forth below:




  
    
      1
      
        
          N
          p
        
        +
        
          N
          s
        
      
    
    ·
    D
    ·
    
      (
      
        
          V
          
            i
            ⁢
            
              
            
            ⁢
            n
          
        
        -
        
          V
          out
        
      
      )
    
  
  =
  
    
      1
      
        N
        s
      
    
    ·
    
      (
      
        1
        -
        D
      
      )
    
    ·
    
      
        V
        out
      
      .
    
  




From the aforementioned equation, the steady-state ratio of the dc component of the output voltage Vout to the input voltage Vin of the power converter can readily be derived as set forth below:





  
    
      V
      out
    
    
      V
      
        i
        ⁢
        
          
        
        ⁢
        n
      
    
  
  =
  
    
      (
      
        
          N
          s
        
        
          N
          p
        
      
      )
    
    ·
    
      
        [
        
          D
          
            (
            
              1
              -
              D
              +
              
                
                  N
                  s
                
                
                  N
                  p
                
              
            
            )
          
        
        ]
      
      .
    
  




Representing the ratio of the output voltage Vout to the input voltage Vin of the power converter by a parameter M, the aforementioned equation can alternatively be represented as:




  D
  =
  
    
      
        
          (
          
            1
            +
            
              
                N
                p
              
              
                N
                s
              
            
          
          )
        
        ·
        
          
            V
            out
          
          
            V
            
              i
              ⁢
              
                
              
              ⁢
              n
            
          
        
      
      
        (
        
          1
          +
          
            
              (
              
                
                  N
                  p
                
                
                  N
                  s
                
              
              )
            
            ·
            
              (
              
                
                  V
                  out
                
                
                  V
                  
                    i
                    ⁢
                    
                      
                    
                    ⁢
                    n
                  
                
              
              )
            
          
        
        )
      
    
    =
    
      
        
          
            (
            
              1
              +
              
                
                  N
                  p
                
                
                  N
                  s
                
              
            
            )
          
          ·
          M
        
        
          (
          
            1
            +
            
              
                (
                
                  
                    N
                    p
                  
                  
                    N
                    s
                  
                
                )
              
              ·
              M
            
          
          )
        
      
      .
    
  




Turning now to FIG. 8, illustrated is a graphical representation of contour plots of the ratio of output voltage to input voltage as a function of the turns ratio of coupled sections of windings and the duty cycle for a power converter constructed according to the principles of the present invention. The graph includes contour plots for various ratios of output voltage to input voltage (i. e., for the parameter M representing a ratio of the output voltage to the input voltage). The graph is representative in that imperfect power converter efficiency as well as delays between the enabled conduction periods for the switches has been ignored. In reality, the duty cycle D may be slightly larger than that shown by the graphical representation provided in FIG. 8.
For a conventional buck power converter, the ratio of output voltage to input voltage is the duty cycle D, resulting in very small duty cycles for low ratios of output voltage to input voltage. As illustrated in FIG. 8, by selecting a suitably high winding turns ratio Np/NS for a tapped inductor, the duty cycle D can be designed to be reasonably large, even for a very low ratio of output voltage to input voltage. For example, for the case of an input voltage of 12 volts and an output voltage of 1.5 volts, by selecting a winding turns ratio Np/NS of about 7:1, the resulting duty cycle is 0.5. For a conventional buck power converter for the same input and output voltages, the duty cycle would be about 0.125, which is significantly smaller than the power converter of the present invention and would result in an inefficient buck power converter design.
Turning now to FIG. 9, illustrated is a graphical representation of waveforms for magnetic flux in one leg of a magnetic device and output current from one active phase leg for a power converter constructed according to the principles of the present invention. The upper waveform provided in FIG. 9 represents the magnetic flux φL in a core leg of a tapped-inductor for a power converter constructed according to the principles of the present invention. In the lower waveform of FIG. 9, the corresponding component of output current io produced by a respective active phase leg of the power converter is illustrated. The aforementioned waveforms are provided for periods associated with a duty cycle D and a complementary duty cycle 1-D of the power converter, and maximum and minimum values of the output current io are also provided therein. The step changes in the output current io result from switching transitions of main switch Qmain and the auxiliary switch Qaux of the power converter (see, for instance, FIG. 1) at the beginning and end of a duty cycle D as evident from the illustration.
From further circuit analysis, it can be readily demonstrated that the dc level of the current Imain associated with the main switch Qmain can be related to the magnetizing current iL and hence, the magnetic flux ΦL in a leg of a magnetic core by the equation:




  
    
      I
      main
    
    =
    
      
        D
        ·
        
          I
          L
        
      
      =
      
        
          
            D
            ·
            
              R
              C
            
          
          ⁢
          
            Φ
            L
          
        
        
          
            N
            p
          
          +
          
            N
            s
          
        
      
    
  
  ,




where D is the steady-state duty cycle of the power converter, and RC is the reluctance of the magnetic path through the magnetic core. The currents Imain, IL represent the average values of the respective currents. In general, upper case letters for variables denote corresponding average values. For example, ΦL represents average value of magnetic flux φL. The corresponding dc level of a current Iaux in the auxiliary switch Qaux is given by the equation:





  
    I
    aux
  
  =
  
    
      (
      
        1
        +
        
          
            N
            p
          
          
            N
            s
          
        
      
      )
    
    ·
    
      (
      
        1
        -
        D
      
      )
    
    ·
    
      
        
          
            R
            C
          
          ·
          
            Φ
            L
          
        
        
          
            N
            p
          
          +
          
            N
            s
          
        
      
      .
    
  




The dc value of the output current lo for the power converter is given by the equation:





  
    I
    o
  
  =
  
    
      (
      
        1
        +
        
          
            
              N
              p
            
            
              N
              s
            
          
          ·
          
            (
            
              1
              -
              D
            
            )
          
        
      
      )
    
    ·
    
      
        
          
            R
            C
          
          ·
          
            Φ
            L
          
        
        
          
            N
            p
          
          +
          
            N
            p
          
        
      
      .
    
  




The dc value ΦL of the magnetic flux φLin the magnetic core is related to the dc output current of the power converter by the equation:




  
    Φ
    L
  
  =
  
    
      
        
          N
          p
        
        +
        
          N
          s
        
      
      
        R
        C
      
    
    ·
    
      1
      
        (
        
          1
          +
          
            
              n
              t
            
            ·
            
              (
              
                1
                -
                D
              
              )
            
          
        
        )
      
    
    ·
    
      
        I
        o
      
      .
    
  




Turning now to FIG. 10, illustrated is a schematic diagram of an embodiment of a power converter (e.g., an n-phase tapped-inductor buck power converter) constructed according to the principles of the present invention, which form a plurality “n” of interleaved active phase legs. The power converter is coupled to a source of electrical power (represented by a battery) for supplying an input voltage Vin at input nodes Ni1, Ni2 thereof, and produces an output voltage Vout at output nodes No1, No2 to power a load represented by a resistor RL. The power converter includes a plurality of coupled pairs of windings having a first section of a tapped winding (with primary turns Np) and a second section of the tapped winding (with secondary turns NS) of tapped inductors Ltp1, . . . , Ltpn. The first and second sections of the tapped windings are preferably coupled on a common magnetic core leg.
Each pair of tapped windings of the tapped inductors Ltp1, . . . , Ltpn is coupled to a corresponding active phase leg formed by a corresponding main switch Qmain1, . . . , Qmainn and a corresponding auxiliary switch Qaux1, . . . , Qauxn. The operation of each active phase leg is substantially similar to that of the single-phase tapped-inductor buck power converter illustrated and described with respect to FIGS. 1, 4, and 5. The power converter preferably includes an output capacitor Cout to filter the output voltage Vout of the power converter. In a preferred embodiment, the timing of the switching periods of each active phase leg of the power converter is successively delayed by a fraction 1/n of a switching period to reduce the output ripple current.
Turning now to FIG. 11, illustrated is a side view of another embodiment of a magnetic device constructed according to the principles of the present invention. The embodiment illustrated in FIG. 11 provides an integrated, two-phase tapped inductor employable, for instance, with the power converter illustrated and described with respect to FIG. 10. The magnetic device has first and second outer legs OL1, OL2 and a center or common leg (designated “CL”) that accommodates a common magnetic flux return path. The magnetic fluxes in the two outer legs are represented as φL1 and φL2. The magnetic fluxes in the outer legs are functionally equivalent to the magnetic flux φL represented in FIGS. 2 and 3. In the integrated magnetics case, the magnetic flux φC in the common return path or the common leg is the sum of the outer leg magnetic fluxes φL1, φL2. The tapped windings Wtp1, Wtp2 (including first and second sections thereof with a tap therebetween and having primary turns Np and secondary turns NS) of the tapped inductors Ltp1, Ltp2 encircle the first and second outer legs OL1, OL2, respectively. The reluctance of the magnetic paths coupling the outer legs OL1, OL2 may be advantageously less than the reluctance of the common magnetic flux return path to distribute the magnetic flux more uniformly in the common magnetic flux return path. The reluctance of the common magnetic flux return path may be suitably increased by including an air gap GP as illustrated therein.
Turning now to FIG. 12, illustrated is a perspective view of another embodiment of a magnetic device constructed according to the principles of the present invention. The magnetic device includes a matrix integrated magnetic device for a power converter (e.g., an n-phase, tapped-inductor buck power converter) having a plurality of outer legs and a common leg to accommodate a common magnetic flux return path constructed according to the principles of the present invention. The matrix integrated magnetic device includes a magnetic core with “n” outer legs (one leg of which is designated “OL”), preferably a single center or common leg (designated “CL”) that accommodates a common magnetic flux return path, and a top plate (designated “TP”). Each outer leg OL preferably accommodates a coupled pair of windings having a first section of a tapped winding (with primary turns Np) and a second section of the tapped winding (with secondary turns NS) of tapped inductors (designated “Ltp1, Ltp2, . . . Ltp3”). In other words, the first and second winding sections of the tapped winding of the tapped inductor may be a coupled pair about the outer legs OL. A plurality of core structures may be used in alternative embodiments to synthesize a matrix integrated magnetic device for a power converter as described herein. The top plate TP thereof substantially completes the magnetic flux return path to the outer legs OL.
Although FIG. 12 illustrates the outer legs OL on one side of the common magnetic flux return path, other geometries are well within the broad scope of the invention such as outer legs distributed on two or more sides of a common leg or outer legs distributed in a substantially circular arrangement around a common leg. An air gap may be included between the top plate TP and the common leg CL to increase the reluctance of the path of the common magnetic flux φC. A substantially nonmagnetic, nonconductive material such as a plastic film may be used to form and dimensionally control the magnetic “air gap” (as it is often referred to), as is well known in the art. Essentially, a magnetic core of the matrix integrated magnetic device integrates an n-phase tapped-inductor arrangement in a single and compact magnetic structure. The winding directions and terminal connections for the first and second sections of the windings of the tapped inductors may be determined according to the dot notation illustrated with respect to FIG. 1.
Other physical configurations for the matrix integrated magnetic device may be used such as provided in the following references, namely, U.S. Pat. No. 6,873,237, entitled Core Structure, to S. Chandrasekaran, et al., issued Mar. 29, 2005, U.S. Patent Publication No. 2003/0198067, entitled Core Structure and Interleaved DC-DC Converter Topology, to J. Sun, et al., published Oct. 23, 2003, U.S. Pat. No. 6,980,077, entitled Composite Magnetic Core for Switch-Mode Power Converters, to S. Chandrasekaran, et al, issued Dec. 27, 2005, U.S. Patent Publication No. 2005/0024179, entitled Extended E Matrix Integrated Magnetics (MIM)Core, to S. Chandrasekaran, et al., published Feb. 3, 2005, and U.S. Patent Publication No. 2006/0038650, entitled Vertical Winding Structures For Planar Magnetic Switched-Mode Power Converters, to V. Mehrotra, et al., published Feb. 23, 2006, all of which are incorporated herein by reference.
Turning now to FIG. 13, illustrated is a reluctance model of a power converter constructed according to the principles of the present invention. The illustrated embodiment provides a reluctance model for an n-phase tapped-inductor buck power converter, wherein RC represents the reluctance of a common leg that accommodates a common magnetic flux return path, assumed without restriction to be linear, and wherein RL represents the reluctance of an outer leg, also assumed without restriction to be linear. The magnetic fluxes are induced by magnetomotive force generators (one of which is designated “MMFG”) that produce magnetomotive forces proportional to the current in a winding times the number of turns therein. The magnetic fluxes in the outer legs are represented by magnetic fluxes φL1, φL1, . . . φLn, and magnetic flux in the common leg is represented by magnetic flux φC. The reluctances of the outer legs are preferably equal as represented by the common symbol RL. In a preferred embodiment, the reluctance RC of the common leg is advantageously greater than the reluctance RL of the outer legs.
The currents iLk represent the magnetizing currents corresponding to the windings in the n outer legs and produce proportional magnetomotive forces in the legs, as represented by the vector equation below, wherein a vector [Fk] represents the magnetomotive force components in each outer leg, and the vector [iLk] represents magnetizing current in each of the outer legs:

[Fk]=(Np+Ns)[iLk], where k ε{1, 2, . . . , n}.

Representing the magnetic flux in each outer leg in an n-phase magnetic device by the a vector [φLk], the following equation can be used to relate the vector [iLk] to the vector [φLk]:




  
    [
    
      
        
          
            i
            
              L
              ⁢
              
                
              
              ⁢
              1
            
          
        
      
      
        
          
            i
            
              L
              ⁢
              
                
              
              ⁢
              2
            
          
        
      
      
        
          ⋮
        
      
      
        
          
            i
            
              L
              ⁢
              
                
              
              ⁢
              n
            
          
        
      
    
    ]
  
  =
  
    
      1
      
        
          N
          p
        
        +
        
          N
          s
        
      
    
    ·
    
      [
      
        
          
            
              
                R
                L
              
              +
              
                R
                C
              
            
          
          
            
              R
              C
            
          
          
            ⋯
          
          
            
              R
              C
            
          
        
        
          
            
              R
              C
            
          
          
            
              
                R
                L
              
              +
              
                R
                C
              
            
          
          
            ⋯
          
          
            
              R
              C
            
          
        
        
          
            ⋮
          
          
            ⋮
          
          
            ⋰
          
          
            ⋮
          
        
        
          
            
              R
              C
            
          
          
            
              R
              C
            
          
          
            ⋯
          
          
            
              
                R
                L
              
              +
              
                R
                C
              
            
          
        
      
      ]
    
    ·
    
      
        [
        
          
            
              
                φ
                
                  L
                  ⁢
                  
                    
                  
                  ⁢
                  1
                
              
            
          
          
            
              
                φ
                
                  L
                  ⁢
                  
                    
                  
                  ⁢
                  2
                
              
            
          
          
            
              ⋮
            
          
          
            
              
                φ
                
                  L
                  ⁢
                  
                    
                  
                  ⁢
                  n
                
              
            
          
        
        ]
      
      .
    
  




From further analysis, the average value ΦC of the magnetic flux φC in the common leg can be estimated using the equation:




  
    
      Φ
      C
    
    =
    
      
        
          (
          
            
              N
              p
            
            +
            
              N
              s
            
          
          )
        
        
          
            (
            
              1
              +
              
                
                  (
                  
                    1
                    -
                    D
                  
                  )
                
                ·
                
                  n
                  t
                
              
            
            )
          
          ·
          
            (
            
              
                R
                L
              
              +
              
                n
                ·
                
                  R
                  C
                
              
            
            )
          
        
      
      ·
      
        I
        o
      
    
  
  ,




where Io is the dc current supplied to a load coupled to a power converter employing the magnetic device. The dc magnetic flux ΦL in an outer leg is given by the equation:





  
    Φ
    L
  
  =
  
    
      
        Φ
        C
      
      n
    
    .
  




In equations above, the quantity nt represents the coupled winding turns ratio Np/NS, and n is the number of active phases.

Turning now to FIG. 14, illustrated is a graphical representation demonstrating ripple cancellation effects for a power converter constructed according to the principles of the present invention. More specifically, the graphical representation provides an exemplary ratio of a peak-to-peak change in magnetic flux in a common leg accommodating a common magnetic return path of a magnetic device (e.g., a tapped inductor embodied in a matrix integrated magnetic device) to a peak-to-peak change in magnetic flux in an outer leg of the magnetic device in accordance with the principles of the invention. The graphical representation provides a ratio of the peak-to-peak change in magnetic flux ΔφC in the common leg of the magnetic device to the peak-to-peak change in magnetic flux ΔφL in one of the outer legs thereof as a function of the duty cycle D. The three cases shown in the graphical representation are for a power converter with two, three, or four active phase legs (i. e., n=2, 3, or 4). For two active phase legs (n=2), the peak-to-peak change in magnetic flux ΔφC in the common leg is zero at about 50% duty cycle, whereas for three active phase legs (n=3), the peak-to-peak change in magnetic flux ΔφC in the common leg is zero at about 33% duty cycle. For the case of four active phase legs (n=4), the peak-to-peak change in the magnetic flux ΔφC in the common leg is zero at about 25% duty cycle, and so on.
At these zero magnetic flux change operating points, the magnetic core loss, which is substantially dependent on high-frequency magnetic flux changes therein, is substantially reduced. In particular, the magnetic core loss in the common leg is substantially zero at these preferred operating points. In addition, since the magnetic flux in the common leg is substantially constant, there is substantially no power loss due to eddy current induction in conductors in close proximity to quick varying fringing magnetic flux. This feature, in a preferred embodiment of a tapped-inductor, multi-phase buck power converter, advantageously accommodates the design of a high-efficiency power conversion device with reduced power loss in portions of the magnetic core material and in windings forming the magnetic device. Thus, a power converter has been introduced with an integrated, multi-phase, tapped-inductor, enabling the design of a modular, scalable power converter that is readily amenable to high output current implementations.
An advantageous feature of a power converter (e.g., an integrated, multi-phase tapped inductor buck power converter) as described herein is the multi-phase pulse-width modulation for inductor current interleaving, resulting in a lower switching ripple level in load current, resulting in a need for smaller output capacitance for a given output voltage ripple specification. Of course, the switch modulation scheme is not limited to the pulse-width modulation method and switch modulation schemes may be used to advantage. Another advantageous feature is higher efficiency due to multiple parallel paths for the load current, resulting in reduced conduction losses in semiconductor devices employed in the system powered by the power converter. The lower switching frequency for a given ripple specification results in reduced switching loss. The power converter can advantageously operate with a higher duty cycle D, which provides improved control bandwidth due to the increase in ripple frequency over the switching frequency by a factor equal to number of interleaving phases. The higher power density can also be advantageously achieved due to smaller magnetic devices and lower output capacitance.
Matrix integrated magnetics is a technique to develop cellular, scalable magnetics assemblies that integrate multiple transformers and/or inductors in a single magnetic core. A core structure (which may advantageously employ planar windings) used in an integrated n-phase tapped-inductor buck power converter consists of multiple winding legs equal to the number of active phase legs, and a shared, air-gapped common leg accommodating a common magnetic flux return path, preferably including a distributed air gap, which may comprise material with different permeability, etc. A top plate may serve as a portion of the magnetic flux return path, or the common magnetic flux return path may be returned through the air. Some of the advantages of a core structure for a matrix integrated magnetic device over a discrete implementation include a low profile due to large common leg cross sectional area, shorter air gap, and improved window utilization for the windings. Another advantage of the core structure is the multi-phase pulse width modulation, resulting in interleaving of winding leg magnetic fluxes in a common leg, leading to lower or substantially zero ripple in the air gap flux. The core structure also provides a more uniform magnetic flux distribution and lower core losses.
It is understood by those skilled in the art that a single or multiple turn center-leg winding may be used with an integrated n-phase tapped-inductor buck power converter to improve filtering, among other advantages. Power topologies employing a center leg winding are described in several of the publications, patents, and patent applications listed herein. It is further understood by those skilled in the art that an integrated n-phase tapped-inductor buck power converter may include any number of phases, and is not limited to a particular number of phases illustrated and described herein.
Turning now to FIGS. 15 to 28, illustrated are views of further embodiments of magnetic devices constructed according to the principles of the present invention. The magnetic devices include magnetic cores and winding geometries formed as tapped inductors. The magnetic devices are applicable in nonisolated and isolated power converter circuit topologies such as, without limitation, buck, boost, half bridge, and full bridge topologies. The magnetic devices are applicable to single phase and multiphase circuit arrangements.
Referring now to FIGS. 15A and 15B, illustrated are elevational and plan views, respectively, of an embodiment of a magnetic core of a magnetic device including multiple (e.g., six) outer legs (one of which is designated “OL” with an area AL). A plurality of windings can be formed around each outer leg OL. The magnetic device also includes a common leg (designated “CL” with an area AC). The magnetic core also includes a top plate (designated “TP”). The magnetic core also includes a gap (designated “GP”) in apertures between the common leg CL and the top plate TP, illustrating that the outer legs OL may be longer than the common leg CL, wherein a gap GP in a magnetic path is formed between the common leg CL and the top plate TP.
Turning now to FIGS. 16A and 16B, illustrated are elevational and plan views, respectively, of an embodiment of a magnetic core of a magnetic device including multiple (e.g., six) outer legs (one of which is designated “OL” with an area AL), around which windings can be formed. A common leg (designated “CL”) is split into two portions, each having an area AC/2. The magnetic core also includes a top plate (designated “TP”). The magnetic core also includes a gap (designated “GP”) in apertures between the common leg CL and the top plate TP, illustrating that the outer legs OL may be longer than the common leg CL, wherein a gap GP in a magnetic path is formed between the common leg CL and the top plate TP.
Turning now to FIGS. 17A and 17B, illustrated are plan and elevational views, respectively, of an embodiment of a magnetic core of a magnetic device including multiple (e.g., six) outer legs (one of which is designated “OL” with an area AL). A plurality of windings can be formed around each outer leg OL. The magnetic device also includes a common leg (designated “CL” with an area AC). The magnetic core also includes a top plate (designated “TP”). The magnetic core also includes a gap (designated “GP”) in apertures between the outer legs OL and the top plate TP, illustrating that the outer legs OL may be longer than the common leg CL, wherein a gap GP in a magnetic path is formed between the common leg CL and the top plate TP.
Turning now to FIG. 18, illustrated are views of an embodiment of a winding of a magnetic device employable about or positioned around an outer leg of the magnetic device. The winding can be constructed from a formed and folded planar conductive sheet such as a copper sheet.
Turning now to FIG. 19, illustrated are views of an embodiment of a magnetic core of a magnetic device including multiple (e.g., six) outer legs and a common leg. The outer legs at either end of the magnetic core are positioned back from the magnetic core ends by a short distance, such as the distance DL illustrated in FIG. 19.
Turning now to FIGS. 20 to 23, illustrated are views of embodiments of magnetic cores of magnetic devices including multiple (e.g., six) outer legs and a common leg. The magnetic cores are suitable for application on a through-hole printed wiring board. Of course, any number of outer legs are well within the broad scope of the present invention.
Turning now to FIGS. 24 to 26, illustrated are views of embodiments of magnetic cores of magnetic devices including multiple (e.g., six) outer legs and a common leg. The magnetic cores are suitable for application on a surface-mount printed wiring board. FIGS. 24B, 25B and 26B illustrate a stepped structure of a magnetic core plate (e.g., a top plate), wherein an outer surface of a magnetic core plate is illustrated with nonuniform height. The stepped structure accommodates forming winding pins as surface-mount leads on an outer portion of the stepped surface.
Turning now to FIGS. 27 and 28, illustrated are views of embodiments of magnetic cores of magnetic devices including multiple (e.g., six) outer legs and a common leg. The magnetic cores are suitable for an application combining surface-mount and through-hole assembly. A stepped structure of a magnetic core plate is also illustrated in FIGS. 27B and 28B.
Thus, power converters and methods of operating and manufacturing the same have been introduced herein. In one embodiment, the power converter includes a magnetic device, the magnetic device including a magnetic core and first and second tapped windings. Each winding of the magnetic device has first and second winding sections, respectively, coupled to the magnetic core, wherein the first and second winding sections are formed from a formed and folded planar conductive sheet. In an advantageous embodiment, the magnetic core includes a common leg, a plurality of outer legs, and a top plate.
In an advantageous embodiment, the formed and folded planar conductive sheet is a copper sheet, and the first and second winding sections are constructed with a formed and folded planar conductive sheet formed with structures usable as pins in a through-hole printed wiring board assembly process. In a further advantageous embodiment, the first and second winding sections are constructed with a formed and folded planar conductive sheet formed with structures usable as pins in a surface-mount printed wiring board assembly process. In a further advantageous embodiment, the first and second winding sections are constructed with a formed and folded planar conductive sheet formed with structures usable as pins in a combined through-hole and surface-mount printed wiring board assembly process.
In an advantageous embodiment, the magnetic core is formed with a stepped structure to accommodate winding pins in a surface-mount printed wiring board assembly process, and the top plate is formed with a stepped structure to accommodate winding pins in a through-hole printed wiring board assembly process.
In an advantageous embodiment, the outer legs are longer than the common leg, and a gap in a magnetic path is formed between the common leg and the top plate. In an advantageous embodiment, the outer legs at either end of the magnetic core are positioned back from magnetic core ends. In an advantageous embodiment, a combined cross-sectional area of the outer legs is substantially equal to a combined cross-sectional area of the common legs.
In an alternative but related embodiment, a power converter comprises a magnetic device. The magnetic device includes a magnetic core with first and second tapped windings having first and second winding sections, respectively, coupled thereto. The magnetic core further comprises a common leg, a plurality of outer legs, and a top plate. The power converter further includes first and second active phase legs, the active phase legs including a main switch and an auxiliary switch, respectively. The main switch of the first and second active phase legs is coupled with an input node of the power converter and the first winding section of the first and second tapped windings, respectively. The auxiliary switch of the first and second active phase legs is coupled to the second winding section of the first and second tapped windings, respectively, and an output node of the power converter. The first and second tapped windings are each coupled to the magnetic core.
In an advantageous embodiment, the first and second winding sections are formed from a formed and folded planar conductive sheet. In an advantageous embodiment, the formed and folded planar conductive sheet is a copper sheet.
In an advantageous embodiment, the first and second winding sections are constructed with a formed and folded planar conductive sheet formed with structures usable as pins in a through-hole printed wiring board assembly process. In an advantageous embodiment, the first and second winding sections are constructed with a formed and folded planar conductive sheet formed with structures usable as pins in a surface-mount printed wiring board assembly process. In an advantageous embodiment, the first and second winding sections are constructed with a formed and folded planar conductive sheet formed with structures usable as pins in a combined through-hole and surface-mount printed wiring board assembly process.
In an advantageous embodiment, the magnetic core is formed with a stepped structure to accommodate winding pins in a surface-mount printed wiring board assembly process. In an advantageous embodiment, the top plate is formed with a stepped structure to accommodate winding pins in a through-hole printed wiring board assembly process. In an advantageous embodiment, the outer legs are longer than the common leg, and wherein a gap in a magnetic path is formed between the common leg and the top plate. In an advantageous embodiment, the outer legs at either end of the magnetic core are positioned back from magnetic core ends. In an advantageous embodiment, a combined cross-sectional area of the outer legs is substantially equal to a combined cross-sectional area of the common legs.
For a better understanding of power electronics, see Principles of Power Electronics, by J. G. Kassakian, M. F. Schlecht and G. C. Verghese, Addison-Wesley (1991). For other references describing applicable topologies or related applications see U.S. Pat. No. 6,549,436, entitled Integrated Magnetic Converter Circuit and Method with Improved Filtering, to J. Sun, issued Apr. 15, 2003, U.S. Pat. No. 6,775,159, entitled Switching Power Converter Circuits Providing Main and Auxiliary Output Voltages,” to K. F. Webb, et al., issued Aug. 10, 2004, U.S. Pat. No. 5,555,494, entitled Magnetically Integrated Full Wave DC to DC Converter, to G. Q. Morris, issued Sep. 10, 1996, and U.S. Pat. No. 6,362,986, entitled Voltage Converter with Coupled Inductive Windings, and Associated Methods, to A. M. Schultz, et al., issued Mar. 26, 2002, U.S. Pat. No. 6,094,038, entitled Buck Converter With Inductive Turn Ratio Optimization,” to Lethellier, issued Jul. 25, 2000, a publication entitled Switching Converters with Wide DC Conversion Range, by D. Maksimović and S. Cuk, IEEE Transaction on Power Electronics, pp. 151-157, January, 1991, a publication entitled Design and Performance Evaluation of Multi-Channel Interleaving Quasi-Square-Wave Buck Voltage Regulator Module, by P. Xu, X. Zhou, P. Wong, K. Yao, and F. C. Lee, HFPC, pp. 82-88, 2000, a publication entitled Static and Dynamic Modeling of Tapped-Inductor DC-to-DC Converter, by M. Rico, J. Uceda, J. Sebastian, and F. Aldana, IEEE PESC, pp. 281-288, 1987, a publication entitled Transformerless DC-to-DC Converters with Large Conversion Ratio, by R. D. Middlebrook, IEEE Power Electronics Transactions, pp. 484-488, 1988, a publication entitled Comparison of Three Topology Candidates for 12V VRM, by Jia Wei, Peng Xu, Ho-Pu Wu, Fred C. Lee, Kaiwei Yao, and MaoYe, IEEE APEC, pp. 245-251, 2001, and a publication entitled Introduction to the Tapped Buck Converter, by Ron Lenk, HFPC 2000 Proceedings, pp. 155-166, 2000, all of which are incorporated herein by reference. For related applications, see U.S. patent application Ser. No. 11/349,637, entitled Power Converter Employing Integrated Magnetics with a Current Multiplier Rectifier and Method of Operating the Same, to Chandrasekaran, et al. filed Feb. 8, 2006, published Aug. 24, 2006 as US-2006/0187684 A1, and U.S. patent application Ser. No. 11/361,914, entitled Control Circuit for a Depletion Mode Switch and Method of Operating the Same, to Rozman, filed Feb. 23, 2006, published Sep. 7, 2006 as US-2006/0198173 A1, which are also incorporated herein by reference.
Those skilled in the art should understand that the previously described embodiments of the power converter (e.g., as a multi-phase, tapped inductor buck power converter) constructed with a matrix integrated magnetics device, and related methods of forming and operating the same, are submitted for illustrative purposes only. In addition, other embodiments capable of providing the advantages as described above are well within the broad scope of the present invention. While the circuit and magnetic designs have been described as providing advantages in the environment of a power converter and, more generally, a power supply, other applications therefor are well within the broad scope of the present invention.
Also, although the present invention and its advantages have been described in detail, it should be understood that various changes, substitutions and alterations can be made herein without departing from the spirit and scope of the invention as defined by the appended claims. For example, many of the topology, circuit, and magnetic designs discussed above can be implemented in different methodologies and replaced by other topology, circuit, and magnetic designs, or a combination thereof, to advantageously form power converters providing reduced magnetic flux changes in a magnetic core of a magnetic device thereof, improved efficiency, reduced size, and reduced costs as described herein.
Moreover, the scope of the present application is not intended to be limited to the particular embodiments of the process, machine, manufacture, composition of matter, means, methods and steps described in the specification. As one of ordinary skill in the art will readily appreciate from the disclosure of the present invention, processes, machines, manufacture, compositions of matter, means, methods, or steps, presently existing or later to be developed, that perform substantially the same function or achieve substantially the same result as the corresponding embodiments described herein may be utilized according to the present invention. Accordingly, the appended claims are intended to include within their scope such processes, machines, manufacture, compositions of matter, means, methods, or steps.

## Claims

1. A power converter, comprising:
a magnetic device, including:
a magnetic core, including:
a common leg,
a plurality of outer legs, and
a top plate; and

first and second tapped windings each having first and second winding sections, respectively, coupled to said magnetic core, wherein said first and second winding sections are formed from a formed and folded planar conductive sheet; and

first and second active phase legs including a main switch and an auxiliary switch, respectively, said main switch of said first and second active phase legs being coupled with an input node of said power converter and said first winding section of said first and second tapped windings, respectively, said auxiliary switch of said first and second active phase legs being coupled to said second winding section of said first and second tapped windings, respectively, and an output node of said power converter.

2. The power converter as recited in claim 1 wherein said formed and folded planar conductive sheet is a copper sheet.

3. The power converter as recited in claim 1 wherein said first and second winding sections are constructed with a formed and folded planar conductive sheet formed with structures usable as pins in a through-hole printed wiring board assembly process.

4. The power converter as recited in claim 1 wherein said first and second winding sections are constructed with a formed and folded planar conductive sheet formed with structures usable as pins in a surface-mount printed wiring board assembly process.

5. The power converter as recited in claim 1 wherein said first and second winding sections are constructed with a formed and folded planar conductive sheet formed with structures usable as pins in a combined through-hole and surface-mount printed wiring board assembly process.

6. The power converter as recited in claim 1 wherein said magnetic core is formed with a stepped structure to accommodate winding pins in a surface-mount printed wiring board assembly process.

7. The power converter as recited in claim 1 wherein said top plate is formed with a stepped structure to accommodate winding pins in a through-hole printed wiring board assembly process.

8. The power converter as recited in claim 1 wherein said outer legs are longer than said common leg, and wherein a gap in a magnetic path is formed between said common leg and said top plate.

9. The power converter as recited in claim 1 wherein said outer legs at either end of said magnetic core are positioned back from magnetic core ends.

10. The power converter as recited in claim 1 wherein a combined cross-sectional area of said outer legs is substantially equal to a combined cross-sectional area of said common legs.

11. The power converter as recited in claim 1 wherein said magnetic device is a matrix integrated magnetic device.

12. The power converter as recited in claim 1 wherein said magnetic device is a tapped inductor.

13. The power converter as recited in claim 1 wherein a reluctance of a magnetic path coupling said first and second winding sections of said first tapped winding is less than a reluctance of a common magnetic flux return path thereof.

14. The power converter as recited in claim 1 wherein said main switch of at least one of said first and second active phase legs is a metal-oxide semiconductor field-effect transistor.

15. The power converter as recited in claim 1 wherein said auxiliary switch of at least one of said first and second active phase legs is selected from the group consisting of:
a metal-oxide semiconductor field-effect transistor; and a diode.

16. The power converter as recited in claim 1 wherein said auxiliary switch of at least one of said first and second active phase legs operates as a synchronous rectifier of said first tapped winding.

17. The power converter as recited in claim 1 wherein said main switch of said first active phase leg is interposed between said first and second winding sections of said first tapped winding.

18. The power converter as recited in claim 1 wherein said main switch of said second active phase leg is interposed between said first and second winding sections of said second tapped winding.

19. The power converter as recited in claim 1 wherein said main and auxiliary switches of at least one of said first and second active phase legs conduct in a substantially complementary manner.

20. The power converter as recited in claim 1 wherein a duty cycle associated with said main and auxiliary switches of said second active phase leg is delayed from said duty cycle associated with said main and auxiliary switches of said first active phase leg substantially inversely proportional to a number of active phase legs of said power converter.