# Patent

## Title
Reverse polarity protection for n-substrate high-side switches

## Abstract

A semiconductor device is disclosed. In accordance with a first aspect of the present invention the device includes a semiconductor chip having a substrate, a first supply terminal electrically coupled to the substrate to provide a first supply potential (VS) and a load current to the substrate, and a second supply terminal operably provided with a second supply potential. A first vertical transistor is integrated in the semiconductor chip and electrically coupled between the supply terminal and an output terminal. The first vertical transistor is configured to provide a current path for the load current to the output terminal in accordance with a control signal, which is provided to a gate electrode of the first vertical transistor.

## Background
TECHNICAL FIELD
The present description relates to circuits providing a reverse polarity protection for semiconductor switches, particularly for a semiconductor device including a plurality of DMOS high side switches integrated in an n-doped substrate.
BACKGROUND
Currently smart power semiconductor switches are used in a broad variety of applications. Not only in automotive applications smart semiconductor switches are increasingly used to replace electro-mechanical relays. Particularly in battery powered systems (e.g., the electronic equipment of a car) a sufficient reverse polarity protection is a requirement. In a typical automotive application the nominal supply voltage is +12 V. A robust electronic device is usually required to withstand a supply voltage up to −16 V (reverse voltage) for at least two minutes.
A multi-channel switching device usually includes one power semiconductor switch (usually a MOSFET) per output channel wherein an electric load is connected to each output channel. Thus each electric load may be switched on and off using the respective semiconductor switch. Assuming an on-resistance of 100 mΩ and a nominal load current of 1 A results in a power loss of 100 mW for each active output channel. Each power semiconductor switch have usually a reverse diode coupled in parallel to the load current path of the semiconductor switch (e.g., the drain-source path in case of a MOSFET). During normal operation this reverse diode is reverse biased and in a blocking state. However, when a negative supply voltage is applied, the reverse diode becomes forward biased and load current may be directed form ground through the load and the reverse diode to the negative supply potential. Assuming a diode forward voltage of at least 0.7 V the resulting power loss in the reverse diode (and thus in the switching device) is 700 mW per output channel, at least seven times more as during normal operation. It goes without saying that such a situation may be hazardous for the switching device and appropriate reverse polarity protection circuitry is required.
Known circuitry providing reverse polarity protection to smart semiconductor switches is comparably complex and requires significant chip space. Thus there is a need for a smart semiconductor switch including an efficient (as far as circuit complexity and chip space requirement are concerned) reverse polarity protection.
SUMMARY OF THE INVENTION
A semiconductor device is disclosed. In accordance with a first aspect of the present invention the device includes a semiconductor chip having a substrate, a first supply terminal electrically coupled to the substrate to provide a first supply potential (VS) and a load current to the substrate, and a second supply terminal operably provided with a second supply potential. A first vertical transistor is integrated in the semiconductor chip and electrically coupled between the supply terminal and an output terminal. The first vertical transistor is configured to provide a current path for the load current to the output terminal in accordance with a control signal, which is provided to a gate electrode of the first vertical transistor.
Furthermore, a control circuitry is integrated in the semiconductor chip and coupled to the first vertical transistor. The control circuitry is configured to generate the control signal to switch the first vertical transistor on and off. The control circuitry comprises a reverse polarity protection circuit.
The reverse polarity protection circuit includes a first MOS transistor coupled in series with a first diode, wherein the MOS transistor and the diode are coupled between the first and the second supply terminals. The reverse polarity protection circuit further includes a first switching circuit that is coupled to the first MOS transistor, electrically connected between the first and the second supply terminals, and configured to activate the MOS transistor when the second supply potential exceeds the first supply potential by more than a given threshold.



BRIEF DESCRIPTION OF THE DRAWINGS
The invention can be better understood with reference to the following drawings and descriptions. The components in the figures are not necessarily to scale, instead emphasis is placed upon illustrating the principles of the invention. Moreover, in the figures, like reference numerals designate corresponding parts. In the drawings:
FIG. 1 illustrates an exemplary switching device with multiple output channels, each channel including one high-side n-channel MOS transistor (exemplary voltage levels are specified for normal operation);
FIG. 2 illustrates the same circuit as FIG. 1 (exemplary voltage levels are specified for reverse polarity operation);
FIG. 3 illustrates a switching device including one exemplary high-side n-channel MOS transistor and a reverse polarity protection circuit in accordance with one example of the invention (exemplary voltage levels are specified for normal operation during active clamping);
FIG. 4 illustrates the same circuit as FIG. 3 (exemplary voltage levels are specified for reverse polarity operation);
FIG. 5 is a cross sectional view through a semiconductor body in which a DMOS power transistor and a CMOS gate is integrated;
FIG. 6 illustrates one detail of the circuit of FIG. 3 including a p-channel MOS transistor (exemplary voltage levels are specified for normal operation);
FIG. 7 illustrates the same detail as FIG. 5 (exemplary voltage levels are specified for reverse polarity operation);
FIG. 8 illustrates the implementation of the p-channel MOS transistor used in the example of FIG. 5 by means of a cross-sectional view of a semiconductor body, the n-doped body zone of the transistor being isolated from the n-doped substrate by a p-doped isolation zone; and
FIGS. 9a and 9b, collectively FIG. 9, include a circuit configured to apply a specific potential to the p-doped isolation zone of the transistor of FIG. 8.



DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS
FIG. 1 illustrates an exemplary multi-channel switching device that includes a plurality of output channels. Each channel includes one high-side semiconductor switch. The examples presented herein are related to n-channel MOS transistors that are used as high-side power semiconductor switches. In particular, vertical power MOS transistors (e.g., DMOS transistors with or without a trench-gate) are considered. In FIG. 1 includes labels indicating the voltage levels for different circuit nodes in case of normal operation (positive supply voltage VS=12 V, ground potential VGND=0 V). FIG. 2 illustrates the same circuit. However, the labels included in FIG. 2 labels indicate the voltage levels for different circuit nodes in case of reverse polarity operation (supply voltage VS=0 V, ground potential VGND=12V).
The exemplary circuit of FIGS. 1 and 2 include a switching device 1 that is supplied with a supply voltage VS at a supply terminal and a corresponding reference potential, further referred to as ground GND, at a ground terminal. The supply voltage may be, for example, provided by an automotive battery. The switching device includes a plurality of n-channel high-side DMOS transistors T1, T2, . . . Tn wherein each one of the transistors associated with a respective output channel. Each transistor T1, T2, . . . , Tn has a load current path (e.g., the drain-source current path in case of a MOS transistor) that (internally) couples a respective output terminal OUT1, OUT2, . . . , OUTn to the supply terminal (potential VS). That is, dependent on the switching state (on or off) of a transistor T1, T2, . . . , Tn a low-resistance current path is provided (via the transistor's load path) from the supply terminal to the output terminal OUT1, OUT2, . . . , OUTn associated with an output channel. Electric loads may be connected between the output terminals OUT1, OUT2, . . . , OUTn and ground. The switching state of a transistor T1, T2, . . . , Tn is set in accordance with a respective control signal (e.g., a gate current or a gate voltage in case of a MOSFET) which is provided, for example, using a gate driver circuit (not shown), wherein a gate driver circuit generates the control signals in accordance with input signals supplied to respective input pins IN1, IN2, . . . INn. One input signal may be provided for each output channel. A logic circuitry 18 may be employed for pre-processing the signals supplied to the input pins IN1, IN2, . . . INn. Typically the logic circuitry generates the control signals supplied to the gate drivers dependent on the signals supplied to the input pins IN1, IN2, . . . INn.
Each transistor T1, T2, . . . , Tn has a reverse diode connected parallel to the transistor's load current path. Usual MOS transistors have an intrinsic reverse diode which always exists due to the internal set-up of the transistor. Other transistors, which do not have an intrinsic reverse diode, may have an external reverse diode to allow free-wheeling when switching inductive loads.
In order to provide a reverse polarity protection of the internal circuitry of the switching device 1 it is known to connect a diode, e.g., a Schottky diode, between the ground terminal of the switching device and the actual ground potential provide by the power supply (e.g., the automotive battery). However, this diode can be replaced by a more sophisticated circuit as will be described later (see also FIG. 9).
The labels indicating the voltage levels present at various circuit nodes refer to voltage levels during normal operation in FIG. 1 and to voltage levels during reverse polarity operation in FIG. 2. During normal operation the ground terminal has a voltage level of 0 V, whereas the supply terminal has a positive voltage level of, e.g., VS=12 V in case of an automotive battery. When an output transistor T1, T2, . . . , Tn is active, the on resistance is low and the voltage drop across the transistor's load path comparably (as compared to the supply voltage) low. In the present example a voltage drop of 100 mV across the output transistors T1, T2, . . . , Tn is assumed so that the voltage at the output terminals OUT1, OUT2, . . . , OUTn is 11.9 V.
During reverse polarity operation (see FIG. 2) the supply voltage of 12 V is applied reversely to the switching device 1. That is, the ground terminal is at 12 V and the supply terminal is at a voltage level of 0 V. As a consequence the reverse diodes DR1, DR2, . . . , DRn become forward bias and thus conductive. The voltage drop across each diode is typically at least 0.7 V (as compared to 0.1 V in the prior case of normal operation) and thus the power losses are at least a factor seven higher than in the case of normal operation. It goes without saying that those high power losses may lead to a thermal destruction of the switching device 1. As a consequence, reverse polarity protection circuitry is required, one example of which is described below with reference to FIGS. 3 and 4.
FIG. 3 illustrates one of the power semiconductor switches included in the switching device 1 shown in FIG. 1. In the present example the power semiconductor switch is implemented as high-side n-channel DMOS transistor T1, which has an intrinsic reverse diode DR1 coupled parallel to the drain-source current path of the transistor T1. As the power transistor T1 is a high-side switch, the supply terminal that receives the supply voltage VS is connected to the drain electrode of the transistor T1. The source electrode of the transistor T1 is connected to the output terminal OUT1 of the respective output channel.
As the power transistor is a vertical n-channel MOS transistor, the supply terminal (and thus also the drain electrode of the power transistor) is electrically connected to the semiconductor substrate in which the power transistor is integrated. Therefore, the substrate has an electric potential VS equal to the potential that is supplied to the respective supply terminal (e.g., 12 V during normal operation, 0V during reverse polarity operation).
The transistors MP1 and MN1 form a CMOS half-bridge (e.g., CMOS inverter) and can be regarded as part of the gate driver circuit GD (i.e., as gate driver output stage). The p-channel MOS transistor MP1 is connected between a floating supply voltage terminal CP, at which a floating supply voltage is applied with respect to the source electrode of the power transistor T1, and the gate electrode of the power transistor T1. The n-channel MOS transistor MN1 is connected between the gate electrode and (e.g., via a resistor R) the source electrode of the power transistor T1. The floating supply voltage may be provided by a bootstrap supply circuit or any other kind of charge pump. Floating power supplies for gate driver circuits are known as such and therefore not further discussed here. In other words, the CMOS half-bridge formed by the transistors MP1 and MN1 is connected between the floating supply voltage terminal CP and a circuit node C which is electrically connected (via the resistor R) to the source electrode of the power transistor T1. The source potential of the power transistor T1 provides the floating reference potential for the floating supply voltage. The inverter output is connected to the gate of the power transistor T1. The resistor R is implemented to provide an ESD protection for protecting the device against electrostatic discharges (short: ESD). Furthermore, the resistor R decouples the source electrode of the power transistor T1 from the floating p-doped well in which the gate driver circuit GD is integrated. This decoupling can be advantageously used during reverse polarity (see below).
During normal operation (i.e., when the battery or any other power supply is correctly connected to the supply terminal and provides a positive supply voltage VS of, e.g., 12 V) the power transistor T1 may be activated (switched on) and deactivated (switched off) either by supplying a positive gate current iG=iON to the gate electrode of the power transistor T1 (thus charging the gate) or, respectively, by draining a negative gate current iG=iOFF from the gate electrode (thus discharging the gate). For charging the gate the p-channel MOS transistor MP1 is activated (e.g., by means of the logic circuitry 18 shown in FIG. 1) while the n-channel MOS transistor MN1 is inactive (switched off). Conversely, for discharging the gate the n-channel MOS transistor MN1 is activated (e.g., by means of the logic circuitry 18 shown in FIG. 1) while the p-channel MOS transistor MP1 is inactive (switched off). It should be noted, that a more sophisticated gate driver circuit may be implemented, e.g., to provide a gate current profile to achieve a specific switching characteristic of the power transistor T1.
As usual, the MOS transistor MN1 of the CMOS inverter has a parasitic npn-type bipolar junction transistor (BJT) Q1 coupled in parallel, wherein the n-doped collector of the BJT Q1 is formed by the drain of the MOS transistor MN1, the base of the BJT Q1 is formed by the p-doped body (bulk) of the MOS transistor MN1, and the n-doped emitter of the of the BJT Q1 is formed by the n-doped substrate, that is connected to the supply voltage VS as discussed above. The implementation of the integrated power transistor T1 and the CMOS gate formed by transistors MN1, MP1 is sketched in FIG. 5 which is a cross-sectional view through (a part of) the semiconductor body in which the switching device 1 is integrated.
FIGS. 3 and 4 further illustrate circuit components which form a reverse polarity protection circuit for protecting the power transistors T1 from the negative effects of reverse polarity as discussed above. A series circuit of an electronic switch SWA and a diode DA may be connected between the ground terminal GND (ground potential VGND) and a circuit node A that is also connected to the gate electrode of the power transistor T1 and the drain electrode of the MOS transistor MN1. The electronic switch SWA is open during normal operation and closed during reverse polarity operation. When closed the electronic switch SWA provides a current path of a defined resistance RA. One exemplary implementation of the switch SWA is discussed later with respect to FIG. 6.
The cathode of the diode DA is coupled with circuit node A whereas the anode of the diode is coupled to ground potential (via the electronic switch SWA). Thus, the diode DA is forward biased during reverse polarity operation as, in this case, the ground terminal is supplied with a high positive supply voltage (e.g., 16 V) while the supply terminal has an electric potential of VS=0V, and the source potential of the power transistor T1 is thus limited to 0.7 V due to the forward biased reverse diode DR1 of the power transistor T1.
For the case of a reversed supply polarity, the series circuit of switch SWA and diode DA may pull the gate of the power transistor T1 to voltage levels high enough to activate the power transistor T1. In this case the power transistor T1 provides a low resistance current path (drain-source current path) that by-passes the reverse diode DR1. As a consequence the voltage drop across the power transistor T1 is about −100 mV (or even lower) whereas the voltage drop (drain-source voltage) would be at least −700 mV if the power transistor T1 were not activated.
The above-mentioned parasitic BJT Q1 may, however, inhibit the activation of the power transistor T1 in case of a reversed supply polarity. During reverse polarity operation an activation (i.e., a forward-biasing of the base-emitter diode) of the BJT may occur and, as a result, the BJT Q1 would become conductive and thus clamp the potential at circuit node A, which is supplied to the gate of the power transistor T1, to about 0 V (i.e., the substrate potential VS during reverse polarity). This clamping of the gate potential to about 0V would inhibit the activation of the power transistor T1 and make the above-described series circuit of diode DA and switch SWA ineffective. So additionally to that series circuit of switch SWA and diode DA for activating the power transistor T1 during reverse polarity operation, further circuit components are required that are able to prevent an activation of the parasitic BJT Q1. For this purpose another electronic switch SWC is used, which is configured to short-circuit the base-emitter diode of the parasitic BJT Q1 and thus to inhibit an activation of the BJT. However, the electronic switch SWC should not be active during normal operation but rather during reverse polarity operation only. Other circuitry than the switch SWC may also appropriate to inhibit the activation of the BJT Q1. As such, the illustrated circuit has to be regarded as an example.
In the example presented in FIGS. 3 and 4 the electronic switch SWC is formed by one or more DMOS transistor cells (forming the DMOS transistor TC). The drain electrodes of the transistor TC and the power transistor T1 are both connected to the supply terminal. This comes naturally when both transistors TC, T1 are integrated in the same manner in the same substrate. To activate (switch on) the transistor TC (and thus the switch SWC) another series circuit of a switch SWB and a diode DB is employed. The series circuit SWB, DB is connected between a circuit node B and the ground terminal GND. The circuit node B is connected to the gate electrode of the transistor TC. During normal operation the switch SWB is open whereas the switch SWB is closed during reverse polarity operation. When closed the electronic switch SWB provides a current path of a defined resistance RB. One exemplary implementation of the switch SWB is discussed later with respect to FIG. 6. While the switch SWB is open (i.e., during normal operation) the transistor TC is switched off, e.g., using the current source XS. The current source XS is coupled between the gate and the source electrode of the transistor TC such that, during normal operation, the gate-source capacitance of the transistor TC is discharged and the transistor TC (and thus the switch SWC) is inactive (switched off).
The function of the circuit of FIGS. 3 and 4 is summarized below and with reference to the exemplary voltage levels given in FIGS. 3 and 4 for normal operation and for a reversed supply polarity, respectively. The voltage levels included in the labels in FIG. 3 represent a state in which the voltage across the inductive load is clamped to a minimum of, e.g., −30 V. A negative voltage can be observed at the output terminal OUT1 when the inductive load (represented by the inductor L1 and the series resistance RL1) is switched off. To limit the negative output voltage in magnitude, the gate of the power transistor T1 (circuit node A) is coupled to the supply terminal via a series circuit of several diodes DA1, DA2, . . . DAn (two, e.g., zener, diodes coupled back-to-back may be sufficient, however) that limits the drain-gate voltage to a maximum, which is 39 V (12 V drain voltage, −27V minimum gate voltage) in the present example. If the output voltage at the output terminal (and thus at the source electrode of the power transistor T1) falls to 30 V, the gate voltage is clamped to the mentioned 27 V and thus the drain-source voltage is high enough to keep the transistor conductive thus allowing the energy stored in the inductor L1 to dissipate in the power switch T1. During this “discharging” of the inductor L1 the output voltage is also clamped to a minimum of, e.g., −30 V. It should be emphasized, however, that the voltage levels indicated in FIG. 3 are merely an illustrative example. The actual voltage levels depend on the actual implementation of the circuit, particularly on the maximum voltage across the diode “chain” DA1, DA2, . . . , DAn.
As mentioned above, the switches SWA, SWB, and SWC are inactive during normal operation (i.e., while the supply polarity is not reversed). Another diode chain DB1, DB2, . . . , DBm may be coupled between the circuit node B and the supply terminal (supply voltage VS) to provide a clamping mechanism to the transistor TC, which forms the switch SWC. The purpose and function of the diode chain DB1, DB2, . . . , DBm is the same as the of the diode chain DA1, DA2, . . . , DAn.
FIG. 4 illustrates the same circuit as FIG. 3. However, the labels indicating the voltage levels refer to a situation with reversed supply polarity. When the supply polarity is reversed, the supply terminal is at VS=0 V and the ground terminal GND is supplied with the positive supply voltage which is 16 V in the present example. As mentioned above, the switches SWA and SWB are switched on during reverse polarity operation. The closed switch SWB pulls the gate of the transistor TC (i.e., the circuit node B) to a voltage level that is high enough (e.g., 5V) to activate the transistor TC. That is, the switch SWC is switched on and thus an activation of the parasitic BJT Q1 is prevented. The output voltage at the output terminal OUT1 is clamped to about 100 mV which is the voltage drop across the switch SWC and thus the maximum base-emitter voltage provided to the BJT Q1 is about 100 mV which is too low to activate the BJT. As an activation of the BJT Q1 is not possible the closed switch SWA pulls the voltage at the circuit node A (gate of the power transistor T1) to a level that is high enough (e.g., 5V) to activate the power transistor T1, thus reducing the voltage drop across the power transistor T1 to about 100 mV. It should be emphasized, however, that the voltage levels indicated in FIG. 4 are merely an illustrative example. The actual voltage levels depend on the actual implementation of the circuit, particularly on the on-resistance of the DMOS transistors T1 and TC.
FIG. 5 illustrates the implementation of the power DMOS transistor T1 as well as the CMOS gate TP1, TN1 in a n-doped substrate. FIG. 5 illustrates a cross-section through the semiconductor body. The cross-section illustrates the power DMOS transistor T1 (at least partly as the transistor T1 is composed of a plurality of transistor cells) in the right side of the illustration and the CMOS half-bridge (transistors MP1 and MN1, see FIG. 3) on the left side of the illustration. The depicted doped zones are not true to scale. Moreover, only those components, which are relevant for the present discussion, are included in the illustration. Oxide layers, some metallization layers, strip lines, etc. have been omitted to allow to concentrate on the relevant parts. In the present example, an epitaxial layer 10′ is disposed (by way of epitaxial deposition) on the silicon substrate 10. The silicon substrate is n-doped, the epitaxial layer 10′ is also made of an n-doped silicon, usually with a lower doping concentration (indicated by the superscript in “n−”). The resulting semiconductor body including the epitaxial layer is often referred to as substrate. It should be noted, that dependent on the manufacturing technology used, the epitaxial layer is optional. The power MOS transistor T1 is implemented as a vertical trench gate transistor. Generally, a vertical transistor is a transistor, in which the load current (i.e., the drain-source current in the case of a MOSFET) is directed from a top surface (where the source electrode is located) of the semiconductor body in a vertical direction through the semiconductor body to a bottom surface (where the drain electrode is located) of the semiconductor body. Transistor cells are formed (and separated by) trenches 46 extending from the top surface into the epitaxial layer 10′ of the semiconductor body. The semiconductor portions between two neighbouring trenches 46 form the transistor cells. In each transistor cell a body zone 41 is formed, e.g., by ion implantation, or diffusion of dopants. In the present example, the body zones 41 are p-doped and extend from the parallel to the top surface of the semiconductor body. Heavily n-doped source zones 44 and heavily p-doped body contact zones 45 are formed (e.g., by way of ion implantation and/or diffusion). The source zones 44 and the body contact zones 45 are extend from the top surface of the semiconductor body into the epitaxial layer 10′ such that they are (in a vertical direction) “enclosed” between the body zones 41 and the top surface of the semiconductor body. In a horizontal direction the source zones 44 are confined by the trenches 46 and the body contact zones 45.
Gate electrodes 42 are formed within the trenches 46 adjacent to the source zones 44 and the body zones 41. Thus, an conductive n-type channel can be established from the source zones 55 through the corresponding body zones 41 to the epitaxial layer 10′ (also referred to as drift zones). The drain of the power transistor cells is formed by the substrate 10. Source electrodes S (usually made of metal) are formed to contact the source zones 44 and the body contact zones 45. The source electrodes S are electrically connected to the output circuit node OUT1 (see also FIG. 3). The gate electrodes 42 are electrically connected to circuit node A (see also FIG. 3).
On the left side of the illustration of FIG. 5 one can see the implementation of the CMOS half-bridge (transistors MP1 and MN1, see also FIG. 3). The NMOS transistor MN1 is integrated in a p-doped well 31 (short: p-well), whereas the PMOS transistor MP1 is integrated in an n-doped well 21 (short: n-well). Both, the n-well 21 and the p-well 31 are formed in the epitaxial layer by way of ion implantation and/or diffusion of dopants, and both are adjacent to each other and separated from the (remaining) epitaxial layer by a p-doped isolation zone 20 (short: p-iso zone), which is a so-called junction isolation zone wherein the isolation is provided by a reverse biased pn-junction between the p-iso zone 20 and the n-doped substrate 10. That is, the n-well 21 and the p-well 31 are enclosed by the p-iso zone 20 and the top surface of the semiconductor body. Within each well 21 and 31 a source zone 24, 34, a corresponding drain zone 23, 33, and a corresponding body contact zone 25, 35 is formed, e.g., by way of ion implantation and/or diffusion of dopants.
The NMOS transistor TN1 is formed within the p-well 31 (also forming the transistor's body zone) by the source zone 34 (n-doped), the drain zone 33 (n-doped), and the gate electrode 32 arranged parallel to the surface of the semiconductor body between the source zone 34 and the drain zone 33. The PMOS transistor TP1 is formed within the n-well 21 (also forming the transistor's body zone) by the source zone 24 (p-doped), the drain zone 23 (p-doped), and the gate electrode 22 arranged parallel to the surface of the semiconductor body between the source zone 24 and the drain zone 23. The body contact zones 25 and 35 are of the same conduction type (p or n) as the respective well 21 and 31, respectively. The drain zones 23 and 33 are electrically connected to circuit node A and thus to the gate electrodes 42 of the power MOSFET T1 (see also FIG. 3). The source zone 34 and the body contact zone 35 are short-circuited and both electrically connected to the source electrode S of the power MOSFET T1 via the resistor R (see also FIG. 3). The source zone 24 and the body contact zone 25 are short-circuited and both electrically connected to the floating supply terminal CP (e.g., coupled to a charge pump) as already described with reference to FIGS. 3 and 4.
The parasitic bipolar junction transistor (BJT) Q1 discussed above with reference to FIGS. 3 and 4 is also sketched in FIG. 5. One can see in FIG. 5 that the emitter of the BJT Q1 is formed by the n-doped epitaxial layer 10′, its collector is formed by the n-doped drain zone 33 (of NMOS transistor MN1), and its base is formed by the p-doped well 31 and the isolation zone 20. As already explained with reference to FIG. 3, the BJT Q1 can electrically connect—when active—the trench gates 42 (i.e., circuit node A) with the substrate 10 and thus with its electrical potential.
FIGS. 6 and 7 illustrate one exemplary implementation of the switch SWA used in the circuit of FIG. 3. The switch SWB may be implemented in an identical manner. In accordance with the present example, the switch SWA is implemented as a p-channel MOS transistor MPA. The source of the transistor MPA is coupled to the ground terminal GND whereas the drain terminal of the transistor MPA is coupled to the circuit node A (via the diode DA). The diode DA blocks any current flowing through the intrinsic reverse diode of the transistor MPA, when the transistor MPA is blocking during normal operation. The gate of the a p-channel MOS transistor MPA is coupled to the ground terminal GND via a resistor R1 and also to the supply terminal (potential VS) via a series circuit of a further diode D3 and a resistor R2. The cathode of the diode D3 is coupled to the supply terminal. Thus the resistors R1 and R2 form a voltage divider with a middle tap connected to the gate of the transistor MPA.
As during normal operation (see FIG. 6) the ground terminal is at 0 V and the supply terminal at a positive supply voltage (e.g., 16 V in the present example) the diode D3 is reverse biased. Thus, the supply voltage drops across the diode D3 whereas the voltage at the middle tap of the voltage divider is at 0 V. As a consequence, the transistor MPA is inactive during normal operation, as the gate-source-voltage is zero. In the present example (and in accordance with the example of FIG. 3) the voltage at the circuit node A is assumed to be −27 V and the voltage drop across the diode DA about 0.7 V, and thus the diode DA has to block about −26.3 V.
During reverse polarity operation (see FIG. 7) the ground terminal is supplied with the positive supply voltage, e.g., 16 V whereas the supply terminal is at 0 V. The diode D3 is forward biased and the middle tap of the voltage divider R1, R2 is pulled up to a voltage low enough (e.g., 14 V) to activate the transistor MPA. In the present example (and in accordance with the example of FIG. 4) the voltage at the circuit node A is assumed to be 5 V and the voltage drop across the diode DA about 0.7 V, and thus the voltage drop across the switch SWA is about 10 V (drain voltage 16 V, source voltage 5.7 V).
FIG. 8 illustrates one exemplary implementation of the p-channel MOS transistor MAP by means of a cross sectional view of the semiconductor body. Accordingly, the MOS transistor MAP may be implemented similar to the transistor MP1 of the CMOS gate mentioned above with regard to FIG. 5. However, the PMOS transistor MAP has to be designed to withstand higher blocking voltages. Accordingly, the PMOS transistor MAP is formed in an n-doped well 11 (short: n-well), which is isolated from the surrounding semiconductor body (or epitaxial layer 10′) by an p-doped isolation zone 20′ similar to the PMOS transistor MP1 illustrated in FIG. 5. A p-doped source zone 14 and a p-doped drain zone 13 as well as an n-doped body contact zone 15 are formed in the n-well 11 by way of ion implantation and/or diffusion of dopants. A gate electrode 12 is arranged parallel to the top surface of the semiconductor body between the source and the drain zone 14, 13. The source zone 14 and the body contact zone 15 are electrically short circuited. In order to avoid an activation of any parasitic bipolar transistor, the electrical potential of the p-isolation zone 20′ should be maintained at or close to 0 V regardless of the operation mode (normal operation or reverse polarity) of the circuit. This task may achieved by the circuit depicted in FIG. 9.
Besides the purpose described above (maintaining the potential of the p-isolation zone 20′ of the transistors MPA and MPB at or close to 0 V, see FIGS. 6 to 8) the circuit of FIG. 9 can also replace the Schottky diode DS depicted in FIG. 1. The circuit of FIG. 9 is connected between the ground terminal GND of the smart switch device 1 and the supply terminal SUP (i.e., the substrate 10), which is provided with the supply potential VS. Two MOS transistors MHV1, MHV2 are connected in series between ground potential VGND present at the ground terminal GND and the supply potential VS present at the substrate (see substrate 10 in FIG. 5). Both transistors MHV1, MHV2 are n-channel MOS transistors having a high blocking voltage. The common circuit node between the two MOS transistors is denoted as internal ground node GNDINT to which the p-isolation zones 20′ (see FIG. 8) are electrically connected. The source electrodes of both transistors are electrically connected to the internal ground node GNDINT whereas the drain electrode of transistor MHV1 is electrically connected to the substrate (and thus to the supply voltage VS) and the drain electrode of the transistor MHV2 is electrically connected to the ground terminal GND. It should be noted that the transistor MHV1 as well as the zener diode DZ coupled between the gate of the transistor MHV1 are used to provide an over-voltage protection and are thus optional for the purpose of reverse polarity protection.
FIG. 9a illustrates exemplary voltages during normal operation (VS=16V, VGND=0V). In this mode of operation, the MOS transistor MHV2 is activated as its gate is charged via the resistor R3 that is connected between the substrate and the gate terminal of the MOS transistor MHV2. That is, the supply potential VS is coupled to the gate of the MOS transistor MHV2 via the resistor R3. As a result, the circuit node GNDINT is almost short-circuited to the (external) ground terminal GND via the MOS transistor MHV2 and the potential at the circuit node GNDINT is approximately 0.1 Volts. The gate voltage of the MOS transistor MHV2 is limited (in the present example to 3 V) by the diode DP, which is coupled between the circuit node GNDINT and the gate electrode of the MOS transistor MHV2. In the present example, the diode DP is a MOS diode which has a forward voltage of about 3 V. A series circuit of several PN-diodes or a reverse biased zener diode may, however, also be used. The labels included in FIG. 9a illustrate voltage labels in the case of a normal operation (VS=16V, VGND=0V).
FIG. 9b illustrates exemplary voltages during reverse polarity (VS=0V, VGND=16V). In this mode of operation, the MOS transistor MHV2 is switched off as the gate is pulled down to 0V via the resistor R3. Further the potential present at the circuit node GNDINT is also pulled down by the resistor R4 which is coupled between the substrate (VS=0V in case of reverse polarity) and the circuit node GNDINT.
Generally, the circuit of FIG. 9 provides an electric potential of approximately 0V to the p-isolation zone(s) 20′ and thus an activation of a parasitic bipolar resistor is avoided regardless of whether the smart switch is operated with normal supply voltage or with reversed supply voltage.
Although exemplary embodiments and its advantages have been described in detail, it should be understood that various changes, substitutions and alterations can be made herein without departing from the spirit and the scope of the invention as defined by the appended claims. With the above range of variations and applications in mind, it should be understood that the present invention is not limited by the foregoing description, nor is it limited by the accompanying drawings. Instead, the present invention is limited only by the following claims and their legal equivalents.
Spatially relative terms such as “under,” “below,” “lower,” “over,” “upper” and the like are used for ease of description to explain the positioning of one element relative to a second element. These terms are intended to encompass different orientations of the device in addition to different orientations than those depicted in the figures. Further, terms such as “first,” “second” and the like, are also used to describe various elements, regions, sections, etc. and are also not intended to be limiting. Like terms refer to like elements throughout the description.
As used herein, the terms “having,” “containing,” “including,” “comprising” and the like are open ended terms that indicate the presence of stated elements or features, but do not preclude additional elements or features. The articles “a,” “an” and “the” are intended to include the plural as well as the singular, unless the context clearly indicates otherwise.

## Claims

1. A semiconductor device comprising:
a semiconductor chip including a substrate;
a first supply terminal electrically coupled to the substrate to provide a first supply potential and a load current to the substrate;
a second supply terminal operably provided with a second supply potential;
a first vertical transistor integrated in the semiconductor chip and electrically coupled between the first supply terminal and an output terminal, the first vertical transistor configured to provide a current path for the load current to the output terminal in accordance with a control signal that is provided to a gate electrode of the first vertical transistor;
a control circuitry integrated in the semiconductor chip and coupled to the first vertical transistor and configured to generate the control signal to switch the first vertical transistor on and off, the control circuitry including a reverse polarity protection circuit that comprises:
a first MOS transistor coupled in series with a first diode, the first MOS transistor and the first diode being coupled between the first and the second supply terminals; and
a first switching circuit coupled to the first MOS transistor and electrically connected between the first and the second supply terminals, the first switching circuit configured to activate the first MOS transistor when the second supply potential exceeds the first supply potential by more than a given threshold, wherein
the control circuitry further comprises a gate driver output stage including an n-channel MOS transistor coupled between the gate electrode of the first vertical transistor and a first circuit node that is electrically connected to the output terminal via a resistor, and
the reverse polarity protection circuit further comprises a second vertical transistor integrated in the semiconductor chip and electrically coupled between the first supply terminal and the first circuit node, the second vertical transistor configured to establish, when activated, a current path between the first supply terminal and the first circuit node.

2. The semiconductor device of claim 1, wherein the first vertical transistor is an n-channel high side transistor and the substrate is an n-doped substrate.

3. The semiconductor device of claim 1,
wherein the reverse polarity protection circuit further comprises:
a second MOS transistor coupled in series with a second diode, the second MOS transistor and the second diode being coupled between the first and the second supply terminals; and
a second switching circuit that is coupled to the second MOS transistor, and is electrically connected between the first and the second supply terminals, the second switching circuit configured to activate the second MOS transistor when the second supply potential exceeds the first supply potential by more than the given threshold.

4. The semiconductor device of claim 3, wherein the first diode is coupled to the gate electrode of the first vertical transistor, and wherein the second diode is coupled to the gate electrode of the second vertical transistor.

5. The semiconductor device of claim 3, wherein the n-channel MOS transistor included in the driver output stage is integrated in a p-doped well arranged in the substrate, the substrate being n-doped.

6. The semiconductor device of claim 3,
wherein the first vertical transistor is an n-channel high side transistor and the substrate is an n-doped substrate composed of a plurality of transistor cells, and
wherein the second vertical transistor is an n-channel transistor composed of one or more transistor cells, the second vertical transistor having fewer transistor cells than the first vertical transistor.

7. The semiconductor device of claim 6,
wherein the substrate is an n-doped silicon substrate including an n-doped well that extends into the substrate from a top surface of the substrate and that is enclosed by a p-doped isolation zone so that the p-doped isolation zone and the n-doped substrate form a junction isolation isolating the n-doped well from the substrate; and
wherein the first MOS transistor and/or the second MOS transistor is a p-channel MOS transistor that is integrated in the n-doped well.

8. The semiconductor device of claim 6,
wherein the driver output stage further includes a p-channel MOS transistor coupled in series to the n-channel MOS transistor thus forming a CMOS half-bridge;
wherein the p-channel MOS transistor is coupled between the gate electrode of the first vertical transistor and a floating supply terminal;
wherein the p-channel MOS transistor included in the driver output stage is integrated in an n-doped well arranged in the substrate; and
wherein the n-doped well is isolated from the substrate by a p-doped isolation zone that forms a junction isolation.

9. The semiconductor device of claim 8, further comprising a circuit coupled between the first supply terminal and the second supply terminal and configured to provide an electric potential to the p-doped isolation zone that is at least approximately equal to ground potential regardless of whether the first supply potential is higher than the second supply potential or vice versa.

10. The semiconductor device of claim 8, further comprising
a further transistor coupled between the second supply terminal and an internal ground node this is electrically connected to the p-doped isolation zone so that the p-doped isolation zone and the internal ground node have substantially the same electric potential;
a further switching circuit coupled to the further transistor and to the first supply terminal and configured to activate the further transistor when the first supply potential exceeds the second supply potential, thus electrically connecting the internal ground node with the second supply terminal; and
a further circuit coupled between the first supply terminal and the internal ground node and configured to pull the electric potential of the internal ground node towards the potential present at the first supply terminal when the second supply potential exceeds the first supply potential.

11. The semiconductor device of claim 10,
wherein the further transistor includes an intrinsic reverse diode, which is coupled parallel to a load current path of the further transistor, and
wherein the further switching circuit includes a further resistor coupled between a gate electrode of the further transistor and the first supply terminal,
such that a current conduction is possible, when the second supply potential exceeds the first supply potential, neither through the further transistor nor through the intrinsic reverse diode.

12. The semiconductor device of claim 10, wherein the further circuit is a pull-down resistor.

13. The semiconductor device of claim 6, wherein the first diode is coupled to the gate electrode of the first vertical transistor, and wherein the second diode is coupled to the gate electrode of the second vertical transistor.

14. The semiconductor device of claim 6, wherein the n-channel MOS transistor included in the driver output stage is integrated in a p-doped well arranged in the substrate, the substrate being n-doped.