Contract J0308024 

University of Oklahoma 

BUREAU OF MINES * UNITED STATES DEPARTMENT OF THE INTERIOR 

Minerals Health and Safety Technology 
The views and conclusions contained in this document 

are those of the authors and should not be interpreted 

as necessarily representing the official policies or recommendations of the Interior Department's Bureau of 

Mines or of the U.S. Government. 
REPORT DOCUMENTATION ='Om 2 3. Rubpant's AccnJon No. 

PAGE OU/EECS/81-1 81/1/USBM 4. t~land jubtttla % Ram Oat* 

Data Security for In-Mine Transmission Final February, 1981 

Report - Part 1 6. 

7. AumoHS) 0. hfformll Omanuatlon Root. No. 

Jon G. Bredeson, Jeffery L. Kohler, Harvinder Singh EECS 1-81 9. hdownlng Orpnkatlon Ham8 and Addms 10. Pmect/Task/Ww(l Untt No. 

School of Electrical Engineering and Computer Science 11. brrCmct(Cl or GmntfG) Ib. University of Oklahoma 

Norman, Oklahoma 73019 to 50308024 

(6) 

12 Spondry Omaniution Harm and Addms It Ty- of R-port L Rrlod Canred 

U.S. Bureau of Mines Final Part I 

Pittsburg Mining and Safety Research Center 14. 11/79 - 12/80- 

4800 Forbes Avenue, Pittsburg, PA 15213 

1% Sup9(ammt.w Nola 

Prepared in cooperation with Ketron, Inc., Wayne, PA 19087 

16. Aktroci (Umk 200 uods) 

A survey of electromagnetic noise reports was performed and a 

summary of the results are given. Industrial concerns were surveyed 

for techniques to minimize the effects of electromagnetic noise and 

are presented. The common types of communication links used in mines 

today are presented. A mathematical development was presented which 

determines the probability of bit errors versus distance for various 

types of communication links designed for in mine use. The probability 

of undetected errors for the protocols of the communication links_isl41 

given together with burst error detection. 

17. Oaumont AnalrJs a. D..cri@on 

Electromagnetic noise, mine communication links, coaxial cable noise, 

twisted pair noise, differential mode noise, common mode noise, data 

security, bit error rates, protocols, probability of undetected error, 

burst errors. h IdMlRen/O#n.Endad Tanns 

Mining Engineering, Electrical Engineering, Communication e. COSATl Fi.(d/Grwp E~ ineerin 

1L Avallabllity Statamant 19. SocurMy Class (This Report) 21. No. of Paga 

unclassified 99 

Release Unlimited a Socurity Class (This Page) 22. Mca 

Z 

TABLE OF CONTENTS 

Section Number 

or Appendix Page 

EXECUTIVE SUMMARY OF FINAL REPORT ................. i 

1 . INTRODUCTION ........................... 1 

2 . ELECTROMAGNETIC NOISE IN MINES .................. 3 

2.1. Introduction ........................ 3 

2.2. SurveyofNoiseMeasurements ................ 3 

2.3. SummaryofNoiseMeasurements ................ 4 

3 . COMMUNICATIONS SYSTEMS IN INDUSTRIAL ENVIRONMENTS ......... 7 

3.1. Introduction ........................ 7 

3.2. Summaryand Conclusions ................... 8 

4 . CHANNELS USED IN MINE COMMLTNICATION SYSTEMS ............ 10 

4.1. Introduction ........................ 10 

5 . THEORETICAL DEVELOPMENT OF NOISE ON MINE COMMLTNICATION 

TRANSMISSIONLINES ...................... 14 

5.1. Introduction ........................ 14 

5.2. Twisted Pair Differential Mode Current ........... 15 

5.3. Twisted Pair Common Mode Current .............. 18 

5.4. Coaxial Cables ....................... 18 

5.5. USBMSpecial Cable ..................... 21 

5.6. SignaltoNoiseRatio .................... 21 

6 . MINE CHANNELERRORPERFORMANCE .................. 25 

6.1. Introduction ........................ 25 

6.2. Conspec Data Communication System .............. 25 

6.3. West Virginia University MACCE Data Communication 

System ........................... 30 
7 . DATA SECURITY ............................ 

7.1. Introduction .......................... 

7.2. Conspec Data Communication System ............... 

7.3. West Virginia MACCE Data Communication System ......... 

8 DETECTION . OF BURST ERRORS ...................... 

8.1. Introduction ........................... 

8.2. Conspec Data Communication System ............... 

8.3. West Virginia University MACCE Data Communication 

System ............................ 

9 . CONCLUSIONSANDRECOMMENDATION .................... 

A . RESPONSE OF A TWISTED PAIR TO INCIDENT UNIFORM AND NONUNIFORM 

ELECTROMAGNETIC FIELDS ...................... 

A.1. Introduction .......................... 

A.2. Line with Parallel Conductors Illuminated by 

a Nonuniform Field ....................... 

A.3. Twisted Pair Illuminated by a Uniform Field .......... 

A.4. Twisted Pair Illuminated by a Field Due to a 

Small Loop ........................... 

B . NOISE IN SHIELDED CABLES EXCITED BY EXTERNAL 

ELECTROMAGNETIC FIELDS ...................... 

B.1. Introduction .......................... 

B . 2 . Shielded Cable Illuminated by an Arbitrary 

Field ............................. 

B.3. Coaxial Cable Illuminated by a Uniform Field .......... 

C . CONVERSION OF THE COMMON MODE CURRENT INTO DIFFERENTIAL MODE CURRENT ...................... 

C.1. Introduction .......................... 

C.2. D.C. Analysis of an Unbalanced Transmission Line ........ 

D . COMMON MODE CURRENT IN A TRANSMISSION LINE HAVING ITS 

DISJ(3INT SEGMENTS ILLUMINATED BY UNCORRELATED UNIFORM FIELDS ... 

D.1. Introduction .......................... 
D.2. DerivationoftheEquation ................... 

E . TYPES OF DIGITAL TRANSMISSION SYSTEMS ................ 

E . 1 . Introduction .......................... 

E.2. BasebandTransmission ..................... 

E.3. DigitalModulationSystems ................... 

E.3.1. Amplitude-Shift Keying (.ASK). ................ 

E . 3.2. Frequency-Shif t Keying (FSK) ................. 

E.3.3. PhaseShiftKeying ...................... 

F . ERRORDETECTIONTECHNIQUES ...................... 

F.1. Introduction .......................... 

F.2. Notation ............................ 

F.3. Principles ofErrorDetection ................. 

F.4. ErrorCalculations ....................... 

F.5. Burst Error Detection ..................... 

G . DATA COMMUNICATION PROTOCOLS ..................... 

G.1. Introduction .......................... 

G.2. Digital Data Communication Message 

Protocol.. DDW ........................ 

G.3. Synchronous Data Link Control.. SDLC .............. 

G.4. OtherProtocols ........................ 

H . KZY WORDS .............................. 

REFERENCES .............................. 
LIST OF FIGURES 

Figure Number Page 

2.1. Electromagnetic Noise Extremes for all Mines 

Collins Report. ..................... 5 

4.1. Common Types of Communication Systems. ........... 13 

5.1. Differential Mode Noise Power vs. Frequency, Mean 

Noise Power, Terminated in Characteris tic 

Impedance 135 ohms. ................... 16 

5.2. Differential Mode Noise Power vs. Frequency, Maximum 

Noise Power, Terminated in Characteristic 

Impedance135 ohms. ................... 

5.3. Common Mode Noise power Versus Frequency, Twisted 

Pair Characteristic Impedance 442 Ohms. ......... 

5.4. Noise Power in Load of Coaxial Cable vs. Frequency, 

LDF4-50 Andrew Corporation, Characteristic 

Impedance 50 ohms, shield grounded at receiving 

end....... .................... 

5.6. Common Mode Power in Load vs. Frequency, USBM 

Special Cable, Shield Grounded on Receiving End ..... 

6.1. Probability of Error vs. Distance (Meters) 

Conspec Accessor Link with USBM Cable .......... 

6.2. Probability of Error vs. Distance (Meters) 

Conspec Long Distance Accessor Link ........... 

6.3. Probability of Error vs. Distance (Meters) 

for Mean Noise Power. West Virginia 

University MACCE Link Center Frequency 1170 Hz. ..... 

6.4. Probability of Error vs. Distance (Neters) 

for Mean Noise Power, West Virginia University MACCE Link, Center Frequency 2125 Hz ........ 

6.5. Probability of Error vs. Distance (Meters) 

for Maximum Noise Power, West Virginia 

MACCE Link Center Frequency 1170 Hz ........... 

6.6 Probability of Error vs. Distance (Meters) 

for Naximum Noise Power, West Virginia 

University MACCE Link, Center Frequency 

2125Hz....... .................. 
Probability of Undetected Error vs. Distance 

(Meters), Conspec Accessor Link with USBM 

Cable ............................ 

Probability of Undetected Error vs. Distance 

(Meters), Conspec Long Distance Accessor 

Link. ........................... 

Probability of Undetected Error vs. Distance 

(Meters) for Mean Noise Power, West Virginia 

University MACCE Link, Center Frequency 1170 Hz. ...... 

Probability of Undetected Error vs. Distance 

(Meters) for Mean noise Power, West Virginia 

University MACCE Link, Center Frequency 2125 Hz. ...... 

Probability of Undetected Error vs. Distance 

(Meters) for Maximum Noise Power, West Virginia 

University MACCE Link, Center Frequency 1170 Hz. ...... 

Probability of Undetected Error vs. Distance 

(leters) for Maximum Noise Power, West Virginia 

University MACCE Link, Center Frequency 2125 Hz. ...... 

Parallel Conductor Line Illuminated by a Nonuniform 

Field. ...................... .... . 

Parallel Conductor Line Illuminated by a Uniform 

Plane Wave ......................... 

Twisted Pair Illuminated by a Uniform Plane Wave. ....... 

Twisted Pair Illuminated by a Field Due to a Small 

CurrentLoop ........................ 

Shielded Cable Geometry .................... 

D.C. Analog Circuit of a Transmission Line. .......... 

D.C. Analog Circuit of a Transmission Line 

After A-Y Transformation .................. 

Wire over a Ground Plane. ................... 

Transmission Line Having Its Disjoint Segments 

Illuminated by Uncorrelated Fields ............. 

Block Diagram of a Digital Data System. (a) Transmitter 

(b) Receiver. ....................... 

Sys tern Model and Wavef oms for Synchronous Baseband 

Digital Data Transmission. (a) Baseband Digital Data 

Transmission. (b) Typical Transmitted Sequence. 

(c) Received Sequence Plus Noise. ............. 

LIST OF TABLES 

Table Number Page 

6.1. Characteristics of Mine Communication Links .......... 27 

7.1. Approximate Maximum Distances in Meters for 

Usable Communication Links ................. 45 

E.1. Probability of Error for Various Communication 

Systems .......................... 81 

F.1. Some Common BCH Codes ..................... 86 
EXECUTIVE SUMMARY 

FINAL REPORT - PART I 

The electromagnetic noise that exists in a mining environment is quite 

severe and was shown to have been reasonably well documented by a number of 

researchers. The noise levels are such that prudent cmunication design requires the careful use of all the electromagnetic noise data available. 

A comprehensive survey was undertaken to determine techniques used in 

various industrial environments to achieve secure communications in the presence of electromagnetic interference. The techniques are rudimentary, but have 

produced reasonably good results. A number of these techniques could be applied to the mining environment. 

The survey of channels used in mine comunication systems indicated that 

16 and 18 gauge twisted pair wire is the most common type. A rugged plastic 

sheath and steel messenger cables are sometimes added to provide additional 

ruggedness. The trolley wire is also extensively used for voice communication. 

All data that is transmitted requires a very narrow bandwidth for present communicat ion systems. 

A comprehensive theoretical development of noise on mine communication 

transmission lines for various types of communication links is presented. The 

noise power at the outputs of the links is shown to be quite high for most 

cases. The noise power greatly affects the data rates on all of these links 

and should be used to evaluate the overall bit error rates. 

Several communication systems intended for use in the mining environment 

are examined. The bit error rate as a function of distance was mathematically 

determined for these systems and plotted. The data communication protocol for 

these communication systems was examined and the probability of an undetected 

error versus distance was mathematically determined and plotted. The burst 

error detection capabilities are analyzed and presented. 

Conclusions and recommendations are presented for different communication 

systems. Significant mathematical developments and background information is 

included in the appendices. 
SECTION I 

INTRODUCTION 

Electromagnetic noise is always present in data transmission systems because noise enters the transmission medium. The transmission medium can be a 

pair of wires, a radio link, coaxial cable, or other types. The addition of 

noise into the data transmission channel greatly affects the data security and 

baud rate of the system. Electromagnetic noise also enters transmission mediums for in-mine data transmission systems. Any proposed data transmission 

system designed for in-mine applications must take into account the effects of 

electromagnetic noise before the problem of data security can be addressed. 

The data communication systems for transmitting sensor, control and data 

information will certainly be expanded at a significant rate in the next several years as automation and remote sensing increases in the mining industry. 

The types of digital data communication channel requirements will necessarily 

be different for sensor, control, and data information. The main requirements 

for each type of channel will be a different baud rate, transmission distance, 

and bit error rate specification. It is anticipated that baud rates up to 40 

kilobaud will be needed. 

Although electrical noise is not a desired physical component of a data 

transmission system, its presence is real. It is not generally associated 

with the transmission mediums, through which it normally gains access to the 

system. During operation, the machinery used in mining creates a wide range 

of many types of intense electromagnetic interference and, therefore, ambient 

electromagnetic interference is a major concern when designing a secure data 

transmission system. With the projected increase of mining by remote control, 

and with the increased emergence of remote environmental and safety monitoring 

systems, there is a definite need to develop and provide secure data transmission hardware and procedures (or protocols) for in-mine systems. A number 

of studies done for the Bureau of Mines have been published which examined the 

electromagnetic noise environment in various types of mines. The electromagnetic fields were measured with several types of antennas in numerous locations 

throughout the mine, as a function of antenna orientation, time of day, and so 

on. Several studies also measured the noise currents and voltages on telephone 

wires and trolley phone wires in various mines. 

The Final Report Part I presents results and findings which are separated 

into sections. The First Section is the introduction and the Second Section will 

consist of a comprehensive review of reports and papers dealing with electromagnetic noise in underground mines. Section Three will be a review of the 

techniques used in various industrial environments which combat electromagnetic 

interference, and an evaluation of implementing these techniques in the mining 

industry. 

A survey of different types of communication transmission systems presently used in mines will be given in Section Four. A mathematical analysis of 

the mechanism that noise enters the transmission medium and plots of noise levels at the ends of communication links will be presented in Section Five. 

SECTION 2 

ELECTROMAGNETIC NOISE IN MINES 

2.1. Introduction 

Several studies done for the Bureau of Mines have been published which examined the electromagnetic noise environment in mines. The electromagnetic 

fields were measured with several types of antennas in numerous locations 

throughout the mine, as a function of antenna orientation, time of day, and so 

on, This noise will be induced into mine communication channels from the electromagnetic environment and will affect the baud rate and bit error rate of 

the communication link. A knowledge of the noise-voltage characteristics that 

exist on mine channels is needed to intelligently design a comunication system. 

2.2. Survey of Noise Measurements 

A number of reports and papers have been published concerning electromagnetic noise in underground mines. Kanda, Bensema, and Adams have obtained considerable data on electromagnetic noise in coal, metal, and non-metal mines 

[1,2,3,4,5]. A portion of this data was also published and summarized by Kanda 

[6]. The time and amplitude statistics illustrated by Kanda are important and 

useful in describing the time-dependent, measured electromagnetic noise in 

mines. They are: 1) Allan variance analyses, 2) interpulse spacing distributions, 3) pulse duration distributions, 4) average crossing rates, and 5) 

amplitude probability distribution. The curves Kanda gives should characterize the noise environment in the mines from which the data was taken and will 

help in the design of reliable communication systems. 

Crary determined that the surface electromagnetic noise can probably be 

best characterized by its sources [7]. These are as follows: 

1. 60 Hz powerline noise and its harmonics. This also includes 

a noise generated on the powerlines by any electrical equipment which may be attached thereto, 

2. Atmospheric noise. The source of this is the distribution of 

thunderstorms over the entire globe at the time in question. 

Signals propagate so well in general at VLF that all noise 

sources will usually contribute significantly to the atmospheric noise at any surface location, with the possible exception of times when an intense local storm is in progress. 

3. VLF-LF transmitters. Their stations operate in the range of 

10 to 40 KHz. They may be significant contributors to the 

noise involved, depending upon the operating frequency and 

locat ion. 

4. "Whistlers and VLF emissions." This group of sources relate 

to solar-geophysical disturbances, dependent upon magnetic 

latitude and the amount of disturbance present. The sources 
was sometimes more than common mode noise which is usually not the case. Cory 

observed that unbalances in the supposedly balanced twisted pair converted a 

lot of the common noise current into differential mode noise current. 

2.3. Summary of Noise Measurements 

The amount of electromagnetic noise whether it is noise current, noise 

voltage, noise magnetic or electric field intensity varies considerably with 

time of day, location in mine, types of equipment operating, and also from 

mine to mine. The noise which would enter a twisted pair will obviously exhibit the same types of variations and to give plots for all these variable 

situations for the different levels of noise would be difficult and not too 

illuminating. A much more reasonable way to proceed which will provide much 

more general results when applied to different mines would be to have a representation of noise for the mean (average) and maximum noise cases. Collins 

Radio Company has summarized and plotted electromagnetic noise for the studies 

conducted by a team from NBS/ITS 1133. This information is plotted in Figure 

2-1 and the mean noise was determined to be approximated by 


SECTION 3 

COMMlJNICATIONS SYSTEMS IN INDUSTRIAL ENYIRONMENTS 

3.1. Introduction 

The techniques used in various industrial environments to achieve secure 

communications in the presence of electromagnetic interference (EMI) were 

examined to determine their applicability for use in the mining industry. 

The review consisted of a brief literature search and extensive meetings with 

technical representatives of various industries. The following types of industries were included in this study: 

Petroleum 

Textile 

Glass 

Manufacturing 

Paper and Pulp 

Chemical 

Automotive 

Steel 

Electric Utility 

During the meetings, the following general information was discussed: 

o The type of communication system used by the different 

industries. 

o General specifications of the hardware used to implement 

the communications system. 

o The physical extent and size of each system. 

o General performance parameters. 

o Research activities performed in support of a new product 

or client who is experiencing EM1 difficulties with a 

given product. 

The following specific information was requested concerning the channels 

used by the cormnunications system: 

o Channel type 

o Bandwidth required by application and bandwidth of channel 

o Tolerable error rate 

o Sources of in-plant EM1 

o Any data on the characteristics of in-plant EM1 

o Type of protocol used 
o Type of error detecting and correcting codes and 

hardware used 

o Specifications of channel (as appropriate for wired 

and wireless links) 

The technical representative of seven (-7) different industries were contacted. Among these individuals, information concerning communications 

systems in over thirty different plants were obtained. In addition, manufacturers of data communications systems were contacted. However, it was impossible to obtain meaningful information from this group due to their concern 

about the proprietary nature of the requested information. The information 

obtained is summarized in the balance of this section. 

3.2. Summary and Conclusions 

All of the communications systems examined were used for process measurement and/or control. These systems have a common denominator in their 

simplicity. It seems that in all cases the emphasis of the systems design was 

on the simple, low cost, and reliable techniques. Almost all of the systems 

in use were analog and utilized a four to twenty milliamp current loop interface. This type of interface was the extent of error detection used by these 

systems. Applications of digital coding or error correcting methods were 

rarely encountered and always consisted of digital parity methods to detect 

errors and initiate retransmission. In cases where extraordinary reliability 

is required, the most prevalent technique is to use redundant transducers and 

data highways. Redundance of data highways is achieved by using multiple 

cables or by sending the signal several times over the same cable. 

The industry standard channel for data transmission is shielded, multiconductor, twisted-pair cable. In many cases, shielded cables are used to 

eliminate all EM1 problems. When questioned on the sources and severity of 

in-plant EMI, it was found that no one had an understanding of this issue. 

It seemed that in all cases, twisted pair was used and if a problem was found 

to exist, the unshielded cable was replaced with a shielded one. Because this 

philosophy has been working so well, there has been no need to gather additional information on EMI, or to resort to a more sophisticated signaling 

technique. The size of the twisted-pair cable is generally determined by 

supplier availability rather than any technical considerations. The only 

deviation from the use of multi-conductor shielded twisted pair occurred in 

electric powered generation plants. There, because of a combination of lowlevel signals and high background-noise, it is necessary to use a cable where 

each twisted pair is shielded and then the entire bundle of cable contains a 

separate shield. This was found adequate for the application. Again, this 

solution resulted from trial and error, rather than an engineering analysis 

of the problem. 

Based on the information obtained in this study, it can be concluded that 

industrial communications system practice for reliability transmitting and 

receiving information in noisy environments is of minimal utility to the 

mining industry. In fact, it is quite surprising to have found that the 

mining industry is really more advanced when it comes to a specific knowledge 

of electromagnetic interference sources and problems in their environment. 
Unfortunately, the industrial solution, i.e., shielded twisted pair whenever 

an EM1 problem is suspected, is not always practical in the mining industry 

for two reasons. First, the extensive lengths of cable required to mining 

would sometimes prohibit the use of the more expensive shielded, twistedpair, multi-conductor cables. Secondly, the maintenance of these cables in 

the harsh mining environment could prove to be difficult. It is encouraging, 

however, to find that secure communications can be implemented in industrial 

environments simply by using shielded cable pairs. It is reasonable to assume 

that manufacturers could supply shielded cables with jacket specially made to 

withstand the mining environment, much as special power cables have been 

developed specifically for mining. 
SECTION 4 

CHANNELS USED IN MINE COMMUNICATION SYSTEMS 

4.1. Introduction 

The purpose of this section was to identify the types of channels being 

used in mine communications systems and to determine the relative performance 

of these channels. During the course of this investigation, technical personnel representing every major coal company in the United States was contacted. The results presented here represent practices in over fifty deep 

coal-mines. 

4.2. Sumary and Conclusions 

Present day communications systems in coal mines consist primarily of 

trolley phones and pager phones. The trolley phone signal is typically frequency modulated with carrier frequencies ranging from 60 kHz to 190 Hz. 

The transmission medium is the trolley line and mine track. The trolley line 

is a very convenient mode of voice communication because it not only provides 

a transmission medium but it also supplies power to the transceiver. The 

typical transceiver is designed to place a 25 watt radio frequency signal onto 

a 20 ohm line, and has a receiver sensitivity of 200 microvolts. On a well 

maintained system, it is possible to have intelligible cormnunications over 

distances of several miles. 

The main disadvantage of using the trolley l i n e a s a communications channel 

is that it is inherently noisy. Sources of noise include rectifier harmonics, 

motor commutation noise, transients from DC contactor opening or closing, 

sparking and arcing resulting from the motion of the trolley pole contact 

shoe on the trolley line, and arcing faults on the DC system. The trolley 

line is also subject to varying impedance resulting from mobile loads (jeeps, 

locomotives, and so forth), movement of pumps, and other DC loads which may 

be placed anywhere along the trolley line. Yet the trolley line still remains 

as the most common mode of voice communication in American coal mines. 

The pager phone is interconnected using two conductor cables. The cable 

type used most frequently consist of two single conductors, PCV insulated, 

number 16 and 18 solid copper wires, which are twisted together. 

Pager phones, which are typically powered by a pair of 12-volt dry cells, 

utilize base-band transmission and are capable of placing a one-or-two-watt 

signal onto a 200 ohm line. Since each phone represents a 4500 ohm load at 

1 kHz, many phones can be placed on one line. Paging is accomplished when the 

initiating phone impresses a 12 or 24 volt DC bias on the line, which activates a speaker amplifier in the other phones. Once comunication has been 

established, conversation is continued using regular telephone handsets with 

a push-to-talk button. A well maintained system can contain up to twenty 

phones and several miles of cable. 

The phone line is inherently less noisy than the trolley line. However, 

the two lines typically are located in the same entry so that there is some 

interaction between the two. For example, the principle harmonic on the 

trolley line (360 Hz) can usually be heard and detected on the phone line. 
The biggest problem with using the phone line for data transmission is 

the condition in which the line is generally maintained. Splices can be 

found at frequent intervals and a "goodf' splice generally consists of twisting the two conductors together with pliers and then applying friction tape. 

After a few weeks the splices become wet, the copper corrodes, and the lines 

short together, or to ground, or become open to various degrees. 

In an attempt to improve the quality of the telephone line (twisted 

pair), some mines are using a number 26 or number 28 solid copper, PVC 

insulated, twisted pair which is enclosed inside a rugged plastic sheath. 

The sheath also contains a steel messenger cable. The transmission capability 

of the line is equivalent to that previously mentioned, and the line is much 

more rugged, thereby reducing the number of splices and other cable damage 

resulting from abuse, roof falls, and so forth. 

Some mines have improved the quality of their transmission by utilizing 

a combination of the two different phone lines. The more expensive steel 

messenger line cable is used wherever the installation will be permanent, and 

the other line is used in the working sections where the line is frequently 

being extended and reduced in length. 

The pager phone and the trolley phone account for approximately 38% of 

all communications system devices in use to date. The pager phone is frequently used to meet the statuatory requirement which requires two-way voice 

communication up to each working section. The remaining 2% of communications 

systems found in deep coal mines consist of experimental devices and monitoring system and remote control systems. 

Typically, the monitoring system include belt monitoring devices for fire, 

slippage, synchronization, and status. A few mines also sense the status of 

circuit breakers and fan parameters. Typically, the twisted pair cable, 

utilized for pager phone communications, is also utilized for the telemetered 

data from remote fans, pumps, and circuit breakers. The requirements of 

these signals are minimal due to the small bandwidth and low data rate. 

Manufacturers installing such monitoring systems have found the maximum single 

frequency on the pager phone line is 7 kHz (typically frequency shift keying 

is used). In a few mines shielded twisted pair can be found, and then a 25 kHz 

signal can be reliably impressed on the line. 

Remote control system, for non-experimental purposes, are rarely found in 

American coal mines. The only application of remote control technology to 

gain acceptance is remote control of continuous mining machines. Both wired, 

through an umbilical cord, and wireless channels are used. The umbilical 

systems utilize a digital asynchronous serial format of rates up to 2000 baud. 

Generally, multiplexing schemes are used to send and receive the data on one 

twisted pair of wires. Relatively simple parity checking methods, and signal 

repetition are employed to increase the data security. The wireless systems 

are similar, except that the digital signal is modulated and transferred at 

a frequency near 450 MHz. 

The greatest problem existing with communication channels in mining 

applications appears to be daily maintenance. The combination of the harsh 

environment along with the lack of knowledge of the miners concerning the 

importance of maintaining channel integrity results in cables which are 
frequently spliced. Errors in data transmission and poor voice quality 

transmissions are always attributed to a faulty channel. When the channel 

is properly repaired the quality of the voice communication is improved and 

data can be transmitted without error. It sh~uld also be noted that the 

existing communication systems in mines do not demand highly secure channels. 
*Useage Code Figure 4.1. Common Types of Communication Systems 

I 

System Type 

1. Pager Phone 

2. Page Phone 

3. Trolley 

Phone 

4. Belt Status 

Indicator 

5. Belt Status 

Indicator 

6. Belt Status 

Indicator 

7. One-\Jay 

Paging 

I Into 

8. Con~munication 

Along 

9. Telephone 

10. Remote Control 

of Machines 

11. Communications 

A Long 

I1 > 90% 

2 751 

3 < 5% 

4 Research Only 

Channel 

Char. 

16-18 Guage 

26-28 Guage 

Noisy 

See "1" and 

"2" Above 

See "1" and 

"2" Above 

See "1" and 

"2" Above 

Roof Bolts 

Used for 

Antenna 

In Umbilical 

Cord 

450 mHz 

Radiax Cable 

(Slotted 

Coaxial) 

Channel Type 

Twisted Pair 

l'wisted Pair 

Trolley Line 

and Rails 

Non-dedicated 

Twisted Pair 

Non-dedicated 

Twisted Pair 

Dedicated 

Twisted Pair 

Wireless 

Wireless 

Coaxial 

Cable 

Wire (Twisted 

Pair Prior) 

Wireless 

Combination 

of Clire and 

ilircl css 

*Usage 

1 

2 

1 

3 

3 

3 

4 

3 

3 

3 

4 

Comments 

Industry Standard 

Tougher Cable 

Industry 

Requires Good Cable 

Maintenance 

Requires Good Cable 

Maintenance 

Not Affected by 

Changes in Voice 

Used to Page People 

Who Aren't Near A 

Phone 

Portable Trancievers 

"Wal kie-Talkie" 

Expensive Cable, 

Slices Require 

Special Tools 

Wireless Trancievers 

Coupled by Radiax 

Cablc 

Frequency and 

Modulation 

0-3kHz Baseband 

0-3kHz Baseband 

60kHz-k90kHz FM 

200Hz Asynchronous 

Baseband 

400kHz FM 

7kHz FSK 

88kHz 

254kHz 

290 FDM Channels 

5.75mHz to 11.75mHz 

2000 Baud 

450 MHz 

450 PMz 

I

Voice 

Voice 

Voice 

Digital Sta

Digital Sta

Digital Sta

Voice 

Voice 

Voice, Moni

Digital Con

Voice 
SECTION 5 

THEORETICAL DEVELOPMENT OF NOISE ON MINE COMMUNICATION TRANSMISSION LINES 

5.1. Introduction 

The electromagnetic noise that exists in a mine will induce a noise voltage or current on a transmission line placed in the mine. The amount of noise 

is a function of the type of transmission line, terminating impedances, electromagnetic field and other factors which are hard to quantify. A number of 

formulas to calculate the noise current and voltages induced on a transmission 

cable from external fields can be found in available literature 116-201. However, the important case of twisted wire pairs does not seem to be covered in 

the current literature. A derivation of the twisted pair case is given in Appendix A with various types of fields. 

The equations for the transmission lines require the electromagnetic 

noise be represented by an electric field intensity (E field) instead of magnetic field intensity (H field). The electromagnetic noise surveys, however, 

have preferred to represent the noise as a magnetic field intensity. The only 

reasonable way to convert from an H field to an E field is to assume the electromagnetic field is a planar wave in free space 1211. The equation relating 

the E field and H field for a planar wave is 

with the E field perpendicular to the H field. Therefore, the electric field 

intensity for mean noise and maximum noise cases will be modified by 

E(dB) = 20 log 377 + H(dB). 

The resultant E fields for electromagnetic noise for the mean and maximum 

cases are 

E(dB) = 173.03 - 32.5 loglOf Mean (Average) 

E(dB) = 210.76 - 33.95 loglOf Maximum 

respectively and are in dB/p volt/meter/ &. 

The electric field intensity will be assumed to be the same in any direction even though noise surveys indicate that the H fields are a function of 

antenna orientation. The rationale for this assumption is that the noise 

fields were measured at certain times and in certain mines and the noise will 

vary. The information from this report will hopefully aid in the design of 

data communication systems in mines and most communication systems must be 

designed for average and worst case noise environments. Therefore using average and maximum noise field values for all orientations will provide meaningful results when the transmission lines have arbitrary orientations. 

The most fundamental quantity to use for noise in communication systems 

is power because signal to noise ratios are power ratios. The noise power 
will be represented in dB's in relation to 1 milliwatt which are abbreviated 

as dBmls. The dBm's will be with respect to load or characteristic impedances 

as is appropriate. 

5.2. Twisted Pair Differential Mode Current 

The twisted pair transmission line is used normally in a differential current mode where any common mode voltage induced equally on both wires will be 

eliminated by the differential input at the receiver. However, the voltage induced along any section of the pair will not necessarily be the same for each 

wire because of phase and magnitude differences of the illuminating electrical 

field. This difference will produce a differential current which flows in the 

terminating impedance at the receiver and will corrupt the incoming signal. 

The portion of the induced voltage which is common to both wires will produce 

a common mode current which can be much higher than the differential mode current. 

Smith has examined the case of a wire pair with no twist embedded in an 

electric field and calculated the differential current flowing in a load resistor for both plane waves and nonuniform fields which are represented by a 

loop or dipole [19]. The twist in a wire pair produces a cancelling of the effect on the differential current because the differences of the induced voltages on each wire will have the opposite sign one pitch length down the wire. 

Unfortunately, no analysis was found for this important case in an extensive 

literature search. An equation was developed for the twisted pair and is given in Appendix A. The equation is difficult to integrate and must be integrated using numerical integration. 

The most important part of any of these equations is the magnitude because this relates to noise power. The differential noise power in dBm is 

plotted for an 18 gauge twisted pair assuming a plane wave excitation in Figure 5.1., assuming a mean noise field. The differential noise power in dBm 

is also plotted for 18 gauge without a twist for a formula developed by Smith 

and given in Appendix A. The im~rovement from no twist to a line with 2 twist 

amounts to 80-90 dR difference. This is an excellent reason tor twisting the 

wire. The noise power In cne'load is plotted for the same case in Figure 5.2, 

but the maximum noise field is used in this case. The oscillations in the 

curves at higher frequencies occurs because the wavelength is close to the 

length of the cable. 

The actual electromagnetic noise field that exists in a mine will obviously not be a plane wave over the entire length of the communication link. 

A much better approximation would be to represent the electromagnetic noise 

by a nonuniform field similar to a field generated by a loop antenna. This 

would more closely approximate fields generated by electric machines in a 

mine. Smith developed formulas for the current at the end of a pair of wires 

with no twist in the vicinity of a loop 1191. A development is given in Appendix A which calculates the current at the end of a twisted pair in the vicinity of a loop. The communication link of 1000 meters would in all probability pass by a number of sources which generate nonuniform fields. Therefore, to make a more reasonable comparison to the plane wave, the number of 

loops along a wire was calculated to give an average field strength approximately the same as for the plane wave. 


The differential noise power in dBm is also plotted for 18 gauge in the 

presence of a nonuniform field in Figure 5.1 and 5.2 for the mean and maximum 

noise field, respectively. The noise power curve for these cases is not as 

smooth as the plane wave cases, however, the noise power for the uniform and 

nonuniform field with a twisted pair is similar. This occurs when the length 

of the pitch is significantly less than a wavelength which is usually the case 

for the frequencies where twisted pair is used. The noise power at the output 

of a twisted pair is more a function of the field strength than if the field 

is uniform or nonuniform. 

5.3. Twisted Pair Common Mode Current 

Any common mode current which flows in a bundle of wires is assumed to be 

divided equally between all wires in the bundle 1191. The common mode current 

in a twisted pair, therefore, should divide equally and 1/2 the total common 

mode current flows in each wire. Cory indicated in his report that due to 

"imbalances" in supposedly "balanced1' transmission links will cause the amount 

of common mode current in each wire to be different and thus effectively produce a differential mode current much higher than anticipated 1121. Therefore, 

the amount of common mode current should also be calculated for the twisted 

pair used in Figure 5.1 and 5.2 under the same conditions. The equation for 

common mode current along a wire bundle is given in Appendix B. The plot of 

common mode current for the same 18 gauge twisted pair as in Figure 5.1 is 

given in Figure 5.3. Appendix C has an analysis for a simple case when a common mode current will not divide equally between both wires in the pair. An 

"imbalance" of one percent rniild then calise a differential mode nnwpr nf 20 dB 

less than the common mode noise Power IlLI. l'hererore, rrom Figure 5.3 tor 

the mean noise case at 1 KHz the common moae noise power is about -15 dBm. 20 

dB less would give -35 dBm differential noise power which is far higher than 

-131 dBm with no twist or -233 dBm with a twist mean differential noise power 

at 1 KHz given in Figure 5.1. This example clearly indicates that considerable care should be taken to insure a twisted pair be closely balanced. 

5.4. Coaxial Cables 

Vance [20] and Smith [19] have shown the illumination of a shielded cable 

by an external electromagnetic field excites a current distribution on the 

outer shield. Since the shield is not a perfect conductor, this current penetrates the shield and produces a voltage distribution along the inside length 

of the cable. This voltage produces a current in the interior load impedances. 

The most effective shield is solid with no holes and of high conductivity. 

The shielding effectiveness for a solid shield increases dramatically when the 

shield thickness is greater than one skin depth [20]. The geometry of a 

shielded cable is given in Appendix B along with appropriate equations. 

A shielded cable that has been used in a mine is a 50 ohm, solid copper 

shielded cable made by Andrew Manufacturing. A plot of the power in the load 

for this cable is given in Figure 5.4. The noise power can be seen to drop 

off rapidly with an increase in frequency when the shield becomes greater 

than one skin depth (occurs at f = 68 KHz). In a cable with a braided shield, 

noise power would not drop off as rapidly due to magnetic and capacitive coupling through holes in the braided shield. Figure 5.4 shows that this coaxial 

cable is a much better communication link when used above 1 MHz. 
c ill 

p :: -70 

cd' 

Ok 

-80 6 

r 

- 

Maximum Power L 

Mean Power 

- 

- 

- 

- .- 

- Length = 1000 m 

- Diameter =.005 m 

.Distance Above Ground = .05 m 

- 

Characteristic Impedance = 442 ohms 

- 

- 

CI 

I 1 

10 100 

Frequency (KHz) 

Figure 5.3. Common Mode Noise Power vs Frequency, Twisted Pair 

Characteristic Impedance 442 Ohms. 
Length 

Diameter 

Wall Thickness 

Dist Above Groun

Characteristic I

Shield Skin Dept

quenc y 

Figure 5.4. Noise Power in Lo

Andrew Corporation Termina

Gro
5.5. USBM Special Cable 

The USBM has contracted to have a special cable built which will carry 

power and signals within the same cables. This cable is illustrated in Figure 

5.5, and has metallized foil shield around each signal conductor plus a metallized foil shield around all conductors in the cable. The current flowing in 

each signal conductor can be found by using a similar equation for a single 

shielded cable with one conductor and by modifying the transfer function. The 

transfer function at the lower frequencies for which this cable is to be used 

is approximately a parallel combination of the DC resistance of the two shields 

with respect to unit length 1201. 

The induced differential current in the two signal conductors will be zero 

because of the surrounding shields. Any differential current which flows will 

be the results of "imbalances" in the transmission line. The common mode power 

in the load resistor when properly terminated is given in Figure 5.6 for this 

cable. The differential mode current caused by "imbalances" in the cable may 

produce a differential mode which is 20-30 dB less than the common mode noise 

power. Again, care should be taken to insure the line is as balanced as possible, when it is used in the differential transmission mode. 

5.6. Signal to Noise Ratio 

The most important parameter to represent the quality of a communication 

channel is the signal to noise ratio at the input to the receiver [22,23]. 

Other parameters that play a significant role will be examined in more depth 

later. The noise power in dBm at the output of the transmission lines 

were calculated for several types of communication links that are used in mine 

environments. The signal to noise ratio in dB's is equal to the signal power 

at the output in dBmls minus the noise power in dBmts at the output. The signal power at the output is equal to the signal power at the input less the 

loss in the transmission link. The noise power at the output in the plots are 

in dBm per Hertz, therefore, to obtain the noise power assuming a bandwidth of 

1000 Hz would require the addition of 30 dB. The attenuation of the twisted 

pair 18 gauge wire is approximately 2 dB per mile. 

A sample calculation for a 1000 meter twisted pair 18 gauge wire to obtain a 20 dB signal to noise ratio at the output is as follows: 

Noise power at output at f = 1 KHz 

(Figure 5.2 Maximum Noise Uniform 

Field with Twist) 

Increase due to 1000 Hz bandwidth 

Total Noise Power 

Signal Output Power for 20 dB SIN 

Transmission loss (1000 meters at 10 KHz) 

Signal input power 

-185 dBm per Hz 

Similar calculations can be carried out for the other communication cables. 

A signal to noise ratio (with no other link impairments) of 20 dB gives a 

reasonably good quality digital communication link. However, the necessary 

SIN ratio is a function of the modulation and demodulation technique used. A 
Figure 5.5. USBM Special Cable. 
Frequency (KHz) 

Figure 5.6. Common Mode Power in Load vs Frequency, USBM Special 

Shield Grounded on Receiving End. 

-25- 

SECTION 6 

MINE CHANNEL ERROR PERFORMANCE 

6.1. Introduction 

Several different digital communication systems have been designed for use 

within a mining environment. Conspec has developed a communication link which 

utilizes a special four wire cable for the main part of the communication system and adapter to transmit on twisted pair for longer transmission links. 

West Virginia University has designed a communication system designated MACCE 

which utilizes twisted pair communication links. These two systems are 

examined in detail to determine mathematically the probability of bit error 

versus distance. [21,22] 

Appendix E contains a summary of the most common digital modulation techniques for low speed data links. The formulas given for the bit probability 

of error are based on the modulation technique, demodulation technique, bandwidth, signal power, and noise power. These formulas assume the noise is white 

and Gaussian with no other impairments in the system 123-251. The impairments 

could include amplitude distortion, phase distortion, crosstalk, impulse noise, 

and etc. These impairments will tend to increase the bit probability of error 

but are difficult to analyze [26]. 

The primary analyses of each communication system will focus on the bit 

probability of error which will take into account the modulation technique, 

type of cable, transmitted power, and noise power. The previous sections mathematically determined the noise power present at the output terminals of various types of communication cables that are placed in the mine. This analysis 

assumed the noise to be represented by a curve of magnetic field intensity 

versus frequency for the mean and maximum noise cases. The mathematical noise 

power calculations required a number of assumptions to be made in order to 

arrive at any answer. The type of electromagnetic waves were assumed to be 

planar or planar over a section of the wire. Appendix D includes a development 

of the common mode noise power for a cable if the electromagnetic noise field 

is approximated by uncorrelated plane waves for only a small portion of the 

wire. This hopefully more accurately represents the fact that a number of different noise sources exist in the mine. The computer programs for common mode 

noise made use of the formulas in Appendix D. 

The assumptions made in almost all cases are conservative in the sense the 

bit probability of error should be less than is mathematically derived, i.s. 

the actual channel performs better. The mathematical development required a 

number of assumptions and simplifications otherwise the analysis would have 

been virtually impossible. 

6.2. Conspec Data Communication System 

Conspec has developed 

mation and control various 

a data communication system which 

units primarily using a special 4 

can collect 

wire cable. 

inf orThis 
pgpp

mately 650 meters. With a signal of 11 volts and mean noise power the maximum 

length jumps to approximately 2,900 meters. 

Conspec uses a long distance accessor trunk CLDA) for transmission over a 

greater distance. The LDA uses 4 wires with a twisted pair for transmission in 

each direction. Frequency shift keying is used with the two frequencies being 

1200 and 2400 Hz with a transmitted signal strength of 0 dBm (1 milliwatt). 

Since this system uses differential transmission it is very important to 

know the differential noise power. The calculations in the previous sections 

together with measured data show that the differential noise power is mainly 

produced by imbalances in the systems which convert a portion of the common 

mode noise power into differential noise power. Cory [11,12,27] has shown that 

the conversion can vary considerably, but with a reasonably constructed twisted 

pair line the differential mode noise should be less than 20 dB below the common mode noise power. Therefore, the computer programs were written which 

mathematically determined the common mode noise power and reduced this by 20 dB 

to arrive at the differential noise power. The computer output was plotted to 

give a probability of error versus distance for the mean and maximum noise case 

which is given in Figure 6.2. The usable distance for a bit error rate of 10-5 

is approximately 900 meters to 8,000 meters for the maximum and mean noise 

power cases, respectively. 
Table 6.1 

CHARACTERIS TICS OF MINE COMMUNICATION LINKS 

Manufacturer 

Conspec 

(Accessor 

Trunk) 

Conspec 

(Long 

Distance 

Accessor 

~runk) 

West 

Virginia 

University 

(MACCE) 

Modulation 

Baseband 

Frequency 

Shift Keying; 

1200 Hz and 

2400 Hz 

Frequency 

Shift Keying; 

1070 Hz, 1270 Hz; 

2025 Hz, 2225 Hz 

Power 

or Voltages 

Transmitted 

0 Volts 

+ 11 Volts 

0 dBm 

(1 Milliwatt) 

0 to -16 

dBm 

Transmission 

Mode 

Single 

Ended 

Differential 

Differential 

Baud Rate 

Bits per 

Second 

4800 

Asynchronous 

Full Duplex 

600 

Asynchronous 

Full Duplex 

300 

Asynchronous 

Full Duplex 

Cable Type 

USBM Cable 

(4 Wires 

transmit, 

receive, 

+24, Ground) 

4 Wires, 

(2 Twisted 

Pairs) 

Twisted 

Pair 

Number 

Bits p

Transmis

22 Bi

22 Bi

Var iab 

Numb er

Maxiqum 

is 204
ri CU mum a h 050 did ri0 I I I I I I I I I C 00030 OCOO I0 I0 I0 'C IS '0 

The MACCE system uses two frequency ranges for transmission in two directions and also has the option of different transmitted signal levels. Therefore, to give an orderly presentation four separate figures are given. Figure 

6.3 gives the probability of error versus distance for the mean noise case with 

the center frequency of 1170 Hz and 0 to -16 dBm transmitted signal power. The 

center frequency is themidpoint of the two transmitted frequencies 1070 Hz and 

is used to determine the noise level. Figure 6.4 gives the probability of 

error versus distance under the same conditions except the center frequency is 

2125 Hz (midpoint of 2025 Hz and 2225 Hz). 

The maximum useful distance for the conditions specified in Figure 6.3 

with Pe z 10-5 range from 3,500 meters to 9,900 meters for -16 dBm and 0 dBm 

transmitted signal power respectively. The similar maximum useful range in 

Figure 6.4 ranges from 4,400 meters to 12,000 meters when the center frequency 

is 2,125 Hz. The reason the distance is greater for 2,125 Hz is the noise 

power falls off as the frequency increases which improves the signal to noise 

ratio (S/N). Small changes in the signal to noise ratio greatly affect the 

probability of error which changes the useful range. 

Figures 6.5 and 6.6 are the same conditions as Figures 6.3 and 6.4, respectively; except the noise is now maximum. The useful range for Pe : 10'5 

varies from 200 meters (350 Pfeters) to 900 meters (1200 Meters) for -16 dBm 

and 0 dBm, respectively, for a center frequency 1170 Hz (2125 Hz) which is 




obtained from Figure 6.5 (Figure 6.6). The figures in paranthesis are for 

Figure 6.6. 

The useful range for the MACCE system is obviously limited by the useful 

distance given for the lower center frequency because both frequency ranges 

are used simultaneously on the twisted pair. The MACCE system for 0 dBm transmitted signal power has a maximum useful range of 9,900 meters (900 meters) 

with a mean (maximum) noise level. This information clearly points out that 

the maximum useful distance is greatly affected by the noise level. 

All of these curves were generated assuming the system twisted pair imbalance caused the differential mode noise to be 20 dB below the common mode 

noise. A better balanced system will obviously increase the maximum useful 

range of the MACCE Communication System and more imbalance will decrease the 

useful range. 
T~ -= time between undetected errors 

'und 

where TB is the time to send one block of data. Also the time to send one 

block is 

TB = NIB where N is the number of bits per block and B is the Baud rate. 

Now let Tund be the average time between undetected errors. Therefore, 

The value chosen for Tund is a matter of engineering judgment as indicated 

before. The value chosen for Tund will be one year in order to make some 

comparison for this report. 

The overall data security of the Conspec and West Virginia University's 

MACCE Data Communication System will be analyzed. The MACCE system gives a 

greater deal of protection because of the error detection scheme. 
7.2. Cons~ec Data Communication Svstem 

The derivation for the probability of an undetected error is given for 

the general case in Appendix P. The Conspec System adds one parity bit for 

each 8 data bits to each asynchronous character that is composed of a start 

bit, 8 data hits, 1 parity bit, and 1 stop bit for a total of 11 bits for 

each asynchronous character. The total message is composed of two asynchronous characters for a total of 22 bits 1211. The probability of an undetected 

error can be found using Appendix F to give 

where p is the bit probability of error. 

e 

The Baud rate is 4800 for the main accessor trunk and Figure 7.1 gives 

the probability of undetected error for the same conditions that bit probability of error was given in Figure 6.1. The average time between undetected 

messages for the main accessor link is 

- - N- - 22 

*und B Pund 4800 *Pund seconds 

Using T = 1 year und 

Note there are 31.5 x lo6 seconds per year. The. maximum useable length of the 

main accessor trunk using the USBM cable would vary from approximately 600 

meters for maximum noise power and 5 volt pulses to approximately 2,700 meters 

for 11 volt pulses and mean noise power. 

Figure 7.2 gives the probability of undetected error for the long distance acces.sor which uses 600 Baud frequency shift keying. The Pund for this 

case is 

The useable distance varies from approximately 750 meters for maximum noise 

power to 7,700 meters for the mean noise power. 

7.3. West Virginia University MACCE Data Communication System 

The MACCE Data Communication System uses frequency shift keying modulation over a twisted pair similar to Western Electrics 103 series data sets 

1221. The MACCE System also specifies the DDCMP protocol which uses a powerful CRC error checking polynomial that can detect single, double, triple, 


The worst case for the MACCE system occurs for the center frequency of 

1170 Hz because of more noise power than at 2125 Hz center frequency. 

Therefore, for the Pund = 2.17 x and the mean noise case gives (Figure 

7.3) a useful range of approximately 4,200 meters at -16 dBm and approximately 11,000 meters at 0 dBm. Similarly the range varies from approximately 

200 meters at -16 dBm to 970 meters at 0 dBm for the maximum noise power 

(Figure 7.5) . 

Table 7.1 gives the approximate maximum useable distance for the comunication links based on the bit probability of error and probability of undetected error which takes the protocol into account. The table gives a clear 

indication of greater effectiveness of the better data security given by the 

CRC polynomial in DDCMP protocol. 



Error Undetected Probability of 

1 

0 

L 

C, CI 

D - = c 2 

OF 
Table 7.1. Approximate Maximum Distances in Meters(Mi1es) for Useable 

Comunicat ion Links. 

Manufacturer 

Conspec 

Accessor 

USBM 

Conspec 

Long Distance 

Accessor 

West Virginia 

MACCE 

Signal Level 

or Power 

5 volts 

11 volts 

0 dBm 

-16 dBM 

- 8 dBm 

0 dBm 

Maximum Distance 

-5 for pe = 10 

Mean 

Noise 

2,100 

(1.3) 

2,900 

(1.8) 

8,000 

(5 .O) 

3,500 

(2.2) 

6,500 

(4.0) 

9,900 

(6 2) 1 

Maximum Distance 

for 1 undetected 

Maximum 

Noise 

650 

(.4) 

800 

(95) 

600 

(-4) 

200 

(01) 

500 

(*3) 

900 

(.6) 

error 

Mean 

Noise 

2,100 

(1.3) 

2,700 

(1.7) 

7,700 

(4.8) 

4,200 

(2.6) 

6,300 

(3.9) 

11,000 

(6.8) 

per year 

Maximum 

Noise 

585 

(.4) 

750 

(*5) 

750 

(05) 

200 

(*I) 

533 

(-3) 

970 

(-6) 
-46- 

SECTION 8 

DETECTION OF BURST ERRORS 

8.1. Introduction 

The question of data security was treated in Section 4 for random errors, 

i.e., those errors which are independent from 1 bit to the next. Burst errors 

are error patterns which occur in groups or bursts. If 1 bit is in error, the 

adjacent bits have a high probability of being in error. The analysis for the 

probability of an undetected burst is considerably different (see Appendix F). 

The CRC polynomial can detect any burst error when the length of the burst is 

equal to or less than the number of redundant bits [31]. Furthermore, a significant number of bursts of length greater than the number of redundant bits 

can be detected if the number of redundant bits is large. 

The probability of a burst being detected given that a burst of a certain 

length has occurred is easy to calculate. The probability of a burst error 

passing undetected through the system is the product of the probability of a 

burst occurring times the probability of being undetected. However, the difficulty is the determination of the probability of a certain length burst 

occurring . 

A sufficient amount of information concerning the occurrence of a burst 

error noise has not been found in the literature. A simplistic approximation 

has been found in the literature. A simplistic approximation has been found 

to burst noise is given in Cory [12J, but is not complete. Therefore, the 

analysis for the different systems will only include the probability of a burst 

error being undetected given the occurrence of the burst error will be covered. 

8.2. Conspec Data Connnunicat ion Svst em 

The Conspec System uses one parity bit for each 8 bits which corresponds 

to a CRC polynomial x + 1 [31]. This system detects all trivial 1 bit burst 

errors, no double burst errors, and only 4 of all burst of length greater than 

2 bits. This same detection system is used for both the normal accessor link 

and the long distance accessor link. 

The burst error protection afforded by the Conspec System is not good. 

Furthermore, intuitively, the probability of burst errors in a mine will be 

high because of the large electrical machines. Hence, the overall chances of 

a burst error passing the system is high. 

8.3. West Virginia Universitv MACCE Data Communication Svstem 

The MACCE system uses a CRC polynomial of degree 16 which insures all 

bursts of length 16 or less are detected, only ~$1~5 = 3.05 x loa5 bursts of 

length 17 are undetected, and (5) l6 = 1.5 x 10-5 of bursts of length more than 

17 are undetected 1311. The burst error protection for the MACCE system is 

very good. 
SECTION 9 

CONCLUSIONS AND RECOMMENDATIONS 

The electromagnetic noise that exists in a mining environment is quite 

severe and has been reasonably well documented by a number of researchers. The 

noise levels are such that prudent communication design requires the careful 

use of all the electromagnetic noise data available. 

A comprehensive survey was undertaken to determine techniques used in various industrial environments to achieve secure communications in the presence 

of electromagnetic interference. Most of the techniques used are rather rudimentary, but have produced reasonably good results. A number of these techniques can be applied to the mining environment. 

A survey of channels used in mine communication systems indicated that 16 

and 18 gauge twisted pair wire is the most common type. A rugged plastic 

sheath and steel messenger cables are sometimes added to provide additional 

ruggedness. The trolley wire is also extensively used for voice communication. 

All data that is transmitted requires a very narrow bandwidth for present communication systems. The ever increasing need for higher data rates will require much more sophisticated cormrmnication systems and hence, a much more 

careful design. 

A comprehensive theoretical development of noise on mine comunication 

transmission lines for various types of communication links was undertaken. 

The noise power at the outputs of the links was quite high for most cases. The 

noise power will greatly affect the data rates on all of these links and must 

be used to evaluate the overall bit error rates. 

A comprehensive development of computer programs was undertaken to determine the bit probability of error for the Conspec Accessor and Long Distance 

Accessor, and the West Virginia University MACCE data communication systems. 

This program included the characteristics of the modulation and demodulation 

techniques and the noise mechanism in which th-e noise entered the communication 

link. A set of plots of bit error rate versus distance for the different systems was presented in Section 6. 

The protocols used by each data communication link specified a error detection scheme which is intended to detect the most probable errors. The probability of an undetected error then becomes an important parameter for data 

security. The probability of undetected error versus distance was plotted and 

presented in Section 7 for the different systems. 

Section 8 contains a brief analysis of burst error detection for the different systems. The analysis was limited only to the probability of a burst 

going undetected given the burst had occurred. 

The Conspec and West Virginia University MACCE data communication systems 

have a reasonable bit prcbability of error versus distance. The main accessor 

trunk of Conspec is driven in a single ended-mode which is more susceptible 

than differential mode to external noise. 
However, the USBM cable contains two shields and the system transmits a 

signal of approximately 35 dBm of power which helps reduce the bit probability 

of error. The long distance accessor and the MCCE system twisted pair cables 

could be replaced with shielded twisted pair and further increase the effective 

distance. 

The West Virginia University WCCE system specifies the DDCm protocol 

which includes a powerful error detecting CRC polynomial. Both Conspec systems 

use only one parity bit for each 8 data bits which gives limited error detecting capabilities. This error detection scheme limits Conspec range and is also 

very susceptible to burst errors. The CRC polynomial in DDC2IP protocol is 

therefore the recommended approach. 

A number of different protocols exist in the data communication industry 

which handle the "handshakingf' between data terminals. International Business 

I%chinels synchronous Data Link Control (SDLC) protocol came on the market approximately the same time as DDCNP. SDLC uses a CRC polynomial which has the 

same random and burst error detecting capabilities as DDCHP, and, therefore, 

SDLC protocol will not give a better or worse error detection than DDCXP. The 

error detection is only a small part of any protocol. Formatting, sequencing, 

acknowledgments, codes, overhead, etc., of each protocol will have a significant affect on data transmission. 

It is therefore recommended that a'complete evaluation of DDCXP, SDLC and 

possibly other protocols for the mining environment be undertalcen. A mathematical 

determination of the probability of burst errors occuring so as to determine burst 

error protection should be examined. Also the effect of shielded twisted pair on 

error rates should be examined. This information is necessary to insure well 

designed in-mine communication link that functions properly and provides for 

excellent data security. This work should be completed before any sophisticated 

and costly new data communication system be installed in a mining environment. 
RESPONSE OF A TWISTED PAIR TO INCIDENT 

UNIFORM All NON UNIFORM ELECTROMAGNETIC FIELDS 

A.1. Introduction 

In this appendix, equations are derived representing the response of a 

twisted pair to incident uniform and non-uniform electromagnetic fields. Although various kinds of non-uniform fields can exist in real world situations, 

this discussion is limited to the specific case of a field produced by a small 

loop. In arriving at the necessary equations, it is convenient to treat the 

case of a line with parallel conductors and then modify the resulting equations to obtain the response for a twisted pair. Section A.2 treats the general case of a parallel conductor line illuminated by any non-uniform electromagnetic field. Section A.3 deals with the response of a twisted pair due to 

the field produced by a small current loop is treated. 

A.2. Line with Parallel Conductors Illuminated by a Non Uniform Field 

Figure A.l shows an isolated two-wire transmission line with parallel conductors having a diameter 'a' and separated by a distance 'b'. The transmission line is contained in the x-z plane with the conductors parallel to the zaxis and the terminations parallel to the x-axis. 21 and 22 are the left hand 

and right-hand terminating impedances respectively. 

The line is illuminated by a non-uniform electromagnetic field having an 

electric field component ~i(x,~,z,w) and a magnetic field component ~i(x,~,z,w). 

The direction of propagation of the field is arbitrary as is shown in the 

Figure A.1. Because of the physical separation of the two conductors the field 

incident on the two conductors will differ in phase. Because of this phase 

difference, a net current would flow through the line and the terminating 

impedances. This current resulting from the phase difference of the incident 

wave is known as the differential mode current and is the only current flowing 

in the terminations of a well balanced transmission line. If the line is 

assumed to be lossless, the differential mode currents 11 and I2 flowing 

through the left-hand and right-hand terminations respectively are given by the 

following equations [19] : 
Figure A.l. Parallel Conductor Line Illuminated 

by a Nonuniform Field 

Direction of propagation. 

t 

Figure A-.2. Parallel Conductpr Line Illuminated 

by a Uniform Plane Wave 
where 

s = the length of the transmission line 

~~~(x,~,z,w) = the component of the incident electric field in the direction of the upper conductor = ~$&,o,z,w). 

i E 2(x,y,z,w) component of the incident electric field in the direction of 

the lower conductor = E~(O,O,Z,W). 

i i Ex, E, = components of the incident electric field in the x and z directions, respectively. 

Zo = characteristic impedance of the line - 

Z = distributed series impedance of the line 

Y = distributed series admittance of the line 

= 2~/h phase constant of the line 

h = wavelength 

f = frequency, hertz 

The first term in equations (A-l) and (A-2) represents the contribution 

along the length of the transmission line and the second and third terms represent the contrihution due to the terminations. 
Further, we take the case of a uniform plane wave which is travelling in 

the x-direction and which is plane polarized in such a manner that all its 

electric field is in the z-direction (as is illustrated in Figure A.2. Since 

the electric field component of such a wave has a zero component in the xdirection, the second and third terms in equations (A-1) and A-2) are zero. 

The resulting equations can be expressed as: 

11 and I2 represent the currents whichwill flow in the terminating impedances in response to the described incident field provided the line is well 

balanced. 

A.3. Twisted Pair Illuminated by a Uniform Field 

Equations (A-5) and (A-6) of the last section can be readily applied to 

obtain the response of a twisted pair to an illuminating plane wave. 

Figure A.3 shows a twisted pair lying along the z-axis. It is illuminated 

by a uniform plane wave travelling in the x-direction and plane polarized in 

the z-direction, as is illustrated in the Figure. Each of the conductors 

and w2 in a twisted pair my be regarded as a helix. The twisted pair is 

assumed to be lying in the z-axis in such a way that the axis of the two 

helical conductors coincide with the z-axis. Consequently the value of the 

x-coordinate (xwl) for wire wl and for known z is given by 

where, 

p = length of one complete twist (as is illustrated in Figure A. 3) . Similarly for wire w2, the x-co-ordinate (.xw2) is given by 

Since in an actual twisted pair, the ratio " is very small, it may be regarded P that ~~(x,y,z,u) isparallel to the conductors on all points on each of the 

conductors. If the phase reference is taken to be z-axis, then, 
where 

i E, = Electric field component in the direction of the conductor at any 

given Goint on the conductor wl. 

i E7 = Electr'ic field component in the direction of the conductor at any L 

given point on the conductor w2. 

where 

i El = Electric field component in the direction of the conductor at 

any given point on the conductor w 1' 

i E2 = Electric field component in the direction of the conductor at 

any given point on the conductor w 2 

from Equations (A-5) and (8-12) we can obtain the following equation: 

b(3 2~rz -t % [2 sin [- cos (-11 COS~(S-z) dz 

0 2 P 

from equations (A-6) and (A-12) we obtain: 

=- bi3 27-r z 

Z, sin[-;;- cos (-) ]sinBz dz 

I1 I2 In the above equations, T and 7 represmt the normalized values 
Y Small loop. 

Figure A--4. Twisted Pair Illuminated by 

Field Due to a Small Current Loop 
the currents flowing though the left-hand and right-hand terminations, respectively, of a well balanced twisted pair. Numerical calculations of the equations (A-13) and (A-14) show that these currents are substantially smaller 

than the corresponding terminal currents that flow in the case of a parallel 

conductor line of section A.1. 

A.4. Twisted Pair Illuminated by a Field Due to a Small Loop 

Figure A.4 shows a twisted pair lying along the z-axis in such a manner 

that the axis of each of its helical conductors coincides with the z-axis. 

The fields radiated by a small current loop can be expressed in the cylindrical co-ordinates by the following equations [19] : 

.sina.e - j 6 (r-R) Radiation Field 

where, 

5 - 

~$(r,$,w) is the electric field component of the radiated field at point - 

(r, 0), in the direction of z-axis. 

i E (R,w) is the electric field component of the radiated field distant R 

from the center of the loop. 

R = distance of the loop from the z axis as shown in the Figure, (A.G). 

Induction field is important in the region wnere Br << 1 and radiation 

field is important in the region where 3r >>I. In the region near z=w (see 

Fig. A.4) both the fields will have comparable contribution. 

As in the previous section the ratio b/p for the helical conductors of 

the twisted pair, is assumed to be small. In such a case direction of each of 

the conductors can be taken as parallel to the z-axis at all points on the 

conductors. 

Further assume b << R. Then, from the known conversions from the cylindrical to rectangular co-ordinations and equations (A-7) and (A-8) we have 

b R + 7 cos (2-rrz 1 p) 
p

Thus, 

hlh2[cos6 h3 - j sin6 h,] Induction (6-23) 

- - I ' 

E~ (R ,o) 1 hih2 [ cos '3 h3 - j sin6 hg] Radiation (A-24) 

(cask? h6 - j sin8 h6) Induction (A-25) ,i 

c ,(z,u) 

(cos6 h6 - j sin6 h ) Radiation (A-26) 6 

Substituting the value of KCu) from equation (A-3) into the equation (.A-1) 

and neglecting the contribution along the terminals we have 

I1 and I2 

represent the normalized curents flowing through the ter- 

(R, w) E~ (R, o) 

minations of a balanced twisted pair due to the induction field. The corresponding values for the currents due to the radiation field can be obtained 

by replacing h2 with h and hg with h4 in equations (A-28) and (A-29). 1 1 
NOISE IN SHIELDED CABLES EXCITED BY EXTEmAL 

ELECTROMAGNETIC FIELDS 

B. 1. Introduction 

In this appendfx, an equation is derived representing the noise current 

inside a shielded cable in response to an illuminating field external to its 

shield. In order to keep the mathematics simple, the case of a coaxial cable 

is considered and the illuminating field is considered to be uniform and 

oriented with its electric field component in the direction of the cable 

length. However, the discussion presented here can be easily extended to 

include any type of field and other types of shielded cables. Section B.2 

discusses the general case of a cable excited by an arbitrary field. In 

section B.3, the equations presented in section B.2 are applied to the specific case of a coaxial cable illuminated by a uniform electromagnetic 

field. A portion of the discussion in this appendix is taken from Smith [19]. 

B.2. Shielded Cable Illuminated by an Arbitrary Field 

The pertinent geometry of a shielded cable excited by an external electromagnetic field is shown in Figure B.l, where 

Electric field component of the incident 

field 

Magnetic field component of the incident 

field 

I (-2 ) Sheath current distribution 

s Length of the cable, meters 

Height of the cable above the ground plane, 

meters 

Z1/2, Z2/2 Terminating impedances of the cable shield 

a Outside diameter of cable, meters 

Characteristic impedance of the cable 

shield 

'a' zb Interior load impedances 

Characteristic impedance of the interior 

of the cable 

Current in the interior load impedance Z b 

x,Y,z Cartesian coordinates 
Direction Propagation 

Ground Plane 

Figure B..l. Shielded Cable Geometry. 
The external electromagnetic field excites a current distribution on 

the outer shield (or sheath). Since the shield is not a perfect conductor, 

this current penetrates along the inside length of the cable. Thfs voltage 

distribution in turn produces a current in the interior load impedances. 

The outer sheath current and the voltage induced along the inside of the 

cable are related by 

where , 

Surface transfer impedance, ohms/meter 

Voltage induced along the inside of the 

cable 

(Further information on cable shielding and surface transfer impedance 

may be found in Reference [32-351 and [17] .) 

If the length of the cable in Figure B.l is taken to be much greater 

than its height above the ground plane (s > > b) then the contribution to 

the sheath current from the ends can be neglected. The load current IL(w) 

in the interior load impedance Zb is obtained by integrating (B.l) over the 

length of the cable. The appropriate transmission line equation is equation 

(A.2) with k(w) = ZtI(z, ) and E; = 0. Thus 

w = 21~f radian frequency 

Bi = 2rr/A. wave number (inside cable) 1 

Although Figure (B.l) depicts a coaxial cable, the approach is valid 

for any shielded cable, whether balanced or unbalanced (e.g. triaxial, 

shielded pair, shielded multiconductor, shielded multicoax etc.). It is 

necessary only to measure the appropriate surface transfer impedance. 

Equation (B.2) is a general equation which can be used to calculate 

the load current spectrum in any shielded cable due to a uniform or nonuniform field of given orientation. Using the transmission line theory, the 

current I (z , w) is first calculated. Subsequent substitution of I (z , w) in 

equation (B.2) and integration yields the value of load current in any 

shielded cable for a given field. 

B.3. Coaxial Cable Illuminated by a Uniform Field 

Consider a uniform plane electromagnetic wave traveling in the y-direction and with its incident electric field component ~~(x,~,z,w) in the zdirection. Thus 
where 

i E is the component of the incident electric field in the z-direction. z 

The current induced in the sheath can then be calculated using the 

transmission line theory and taking the effect of ground into consideration. 

The current induced in the sheath is a common mode current and an expr2ssion 

for it has been derived [19] : 

where, 

D = (Z Z .+ Z Z )cos$s + j(~i + Z 7 )sinBs 

01 02 l"2 

Assuming that the cable is terminated in its characteristic impedances 

Substituting (B-3) in (B-2) and integrating, yields the 2:cpression for I (9). L 

Thus 
-Z1Z2 sin 6s 4- j (2 Z + Z Z + Z Z cosBs - Z1Z2 cosBs) 02 12 ol Jj 1(B-4) 

The above equation represents the noise current flowing in a coaxial 

cable due to a uniform plane electromagnetic field with its electric field 

component in the z-direction, assuming the cable is terminated in matching 

impedances. 
C.2. D.C. Analysis of an Unbalanced Transmission Line 

Figure C.1 shows the d.c. analog circuit of a transmission line. 

The resistance R represents the d.c. resistance of each of the conductors 

of the transmission line and r is a lumped resistor inserted in one of its 

conductors. Typically, the resistor r may be a contact resistor of a 

soldered joint in one of the conductors or it may be any other type of 

resistance, causing imbalance. Z1 and Z2 represent the left hand and 

right hand terminating impedances respectively. Impedances Z represent 

the impedances'of the transmission line to ground. In an actual line they 

may be mostly capacitive but for the purposes of this discussion they are 

considered as resistive. The d.c. voltage source of Figure C.l produces 

currents I1 and 12. These currents are analogous to the common mode currents induced in a practical transmission line due to excitation by external 

electromagnetic fields. Because of the lumped resistor in one of the 

conductors, currents I1 and I2 differ in magnitude slightly. This difference results in a net current flow in impedances Z1 and Z2. This current 

flow can be readily calculated by transforming the A-configuration BDG and 

ACF of Figure C.l to Y-configurations represented by B1N2, D1N2, GIN2 and 

AIN1, C'N1, FIN1, in Figure C.2. In Figure C.2, 
Figure C.1. D.C. Analog Circuit of a Transmission Line 

Figure C.2. D.C. Analog Circuit of a Transmission Line 

after A-Y Transformation 
By writing the loop equation in loop BGD and the node equations at 

nodes By G and D, the receiving end current i can be derived. Thus, R 

Substitution of I1 and I from (C-6) and (C-7) yields: 2 

Under the assumptions 

Z>>R+r 
and 

equation (C-9) reduces to: 

The result shows that the current iR resulting from the imbalance is 

dependent upon the ratio ~/r and its magnitude can vary from zero to V/Z. 

Results obtained from the exact analysis can be expected to be similar. 

\ 
-68- 

APPENDIX D 

COMMON MODE CURRENT IN A TRMSIYISSION LINE HAYING ITS DISJOINT 

S EMENTS ILLUNINATED BY LNCORRELATED UNIFOKY FIELDS 

3.1. Introduction 

It was discussed in previous sections, that in practical transmission 

lines, the value of the Common ?lode Current induced due to external electromagnetic fields can be large. If a mall fraction of this current splits up 

into the differential mode current due to the system imbalance, it can cause 

serious noise problems. Therefore it is of concern to be able to calculate 

the value of the induced common mode current, fairly accurately. An expression for the calculation of the common mode current was given in Appendix B, 

equation (.B-3) for a single wire transmission line, when such a line is in 

the proximity of a perfectly conducting ground plane, and is illuminated by 

a uniform field along its entire length. However in a mining environment, it 

is not likely that a transmission line will be illuminated by the sane uniform field throughout its length. It is reasonable to assume that different 

sections of a line will be illuminated by either unifom or nonuniform fields. 

In this appendix, an equation is derived for the comon mode current in a 

line whose various sections are illuminated by different fields. If the segments are chosen small enough, any field may be considered uniform within a 

segment length. Therefore the derivation of the equation in Section A.2., is 

limited to the case where the'various segments of a line are illuminated by 

uniform fields only. A portion of the discussion in the following section 

is taken from Smith [I]. 

D.2. Derivation of the Equation 

Figure D-l(a) shows a single wire transmissian line. It is of length 

S and terminated at both ends to a perfectly conducting ground plane. The 

terminating impedances are Z1/2 and Z2/2 as is shown in the Figure. The wire 

is of diameter 'a' and the distance between the wire and the ground is chosen 

to be b/2. It is illuminated by a uniform plane wave traveling in the zdirection and having its electric field component oriented in the x-direction. 

An expression for the comon mode current can be derived using the transmission line equations and using the image theory to take the effect of ground 

into consideration. Figure D.l(b) shows the ground plane re~l~ced by an 

image of the wire and its terminations. The necessary inage field is also 

shown in the same figure. Assuming the line to be lossless, the expression 

for the common mode current I(z+,w) at any point zk along the line can be 

expressed in the integral form as [19]. 
Direction of Propagation 

a) Physical Representation 

a 

1 

b) Image Problsm 

z 

0 

Plane 

i * 

Figure D .l. Nire Qver a Ground Plane. 

b / 2 

\ f 

z=s z 

Ground 
where, 

Zo = Z/Y = characteristic impedance of equivalent two-wire line (-i.e. 

the wire and its image) 

Z = distributed series impedance of the equivalent two-wire line. 

distributed shunt admit t anc e equivalent two-wire line. 

i: E (x,y,w) = incident field in z-direction along the wir2 

Z 

3- = 2.rr/X, phase constant 

f = frequency, hertz 

Equation (D-1) can be easily modified to obtain the required equation for 

the common mode current in a transmission line whose different segments are 

illuminated by different fields. Figure D-2 shows a short segment of length 

Pz (S/AZ is an integer) illuminated by some field E;(X,~,U). It is assumed 

that the segnent is short, so that E:(X,~,U) can beLconsidered uniform.along- 1 the segment length. Further, let u/Az = rn and r/Az = m - 1, and let E_(z,y, 

w) be defined by the equation below: 

elsewhere 

i 

For z* 2 u, substituting the value of E z from (9-2) in (PI): 

YZ cosb(s-z) + jZ2 sinb(s-z)] 

'-0 i 

I ,(z*,w) = 2E I* 

m ZoDB z 

{lo(.sinlu - sin?r) - j Z 1 (cosiu - cosl~) % 

J 

(9-3) 

where 

I&(z*,w) = common mode current at z" (for z" 2 u) due :he illumination 

of mth. segment. 
Similarly for z* 5 r, substituting (D-2) in (D-l) : 

Zo king(.s-Tl - sin8 (s-r)] + j Z2 rcosg - (s-u) - cosg ($-?)I 

0-4) 

where 

I,(.z*,a) = common mode current at z* (for z* - < r) due to the illumination of mth segment. 

Combining 0-3) and 0-4) and substituting z* = z. 

Substituting r = (m-1)Az and u = mAz: 

I,(z,o) represents the common mode current at any z along the length of 

the line, due to the illumination of the mth segment of the line. 

The above equation (D-6) can be used to calculate the comon mode current 

in the general case, when some or all of the n segments of the line may be 

illuminated by separate fields (-Figure De2), If transfer function T is 

m defined as: 

then 

due 

... 

I(-z,u) is the total comon mode current at any point 

t? the illumination of its segments by uniform fields 

El etc., when some of the El are nonzero. 

zn ' zm 

z.along the l&ne, 1 EZ1, ~$2 . . .El 

zm 
APPENDLX E 

TYPES OF DIGITAL TRANSHISSION SYSTMS 

E.1. Introduction 

The purpose of this Appendix is to consider various systems for the transmission of digital data and their relative performances. Before beginning, however, consider the block diagram of a digital data transmission system, shown 

in Figure E.1. The focus of our attention will be on the portion of the system 

between the blocks labeled encoder and decoder. In order to gain a better perspective of the overall problem of digital data transmission we will briefly 

discuss the operations performed by the blocks shown as dashed lines. 

While many sources result in message signals which are inherently digital, 

such as teletypewriter and computer signals, it is often advantageous to represent analog signals in digital form (referred to as analog-to-digital conversion) for transmission and then convert them back to analog form upon reception 

(referred to as digital-to-analog conversion). Pulse code modulation (PCX), is 

an example of a modulation technique which can be employed to transmit analog 

messages in digital form. The signal-to-noise ratio performance characteristics of a PCN system, show one advantage of this syst2m to be the option of 

exchanging bandwidth for signal-to-noise rat io improvement [23,24,25] . 

/ Source !J ha'0g/ digital W ~n~o~sr l-4 yad*:a:.; io ' ~taind 

Op clonal 

I 

asen; if Carrier 

source is 

digital 

I I 

Carter re£ Clock Optional Absent if 

(,cot.,erent (sync\. sink (user) 

SYS:E~) SYS t en) needs digi- tal outpu: 

Figure E.1. Slock Diagram of a Dizital Data Systea. 

(a) Transiniccer . (5) 3eceiver. 

The conversion of an znalog signal to a digital signal is becoming much 

nore common because once the signzl has been accurately converted to a digital 
value, the accuracy of the signal is very easy to maintain in transmission. 

Furthermore, the accuracy is determined by the number of bits in the digital 

signal. 

Regardless of whether a source is purely digital or an analog source that 

has been converted to digital, it may be advantageous to add or remove redundant digits to the digital signal. Such procedures, referred to as coding, 

are performed by the encoder-decoder blocks of Figure E.l and will be considered later [23,24,25]. 

With these preliminary considerations over, we now return to the basic 

system in Figure E.1 shown as the blocks with solid lines. If the digital signals at the modulator input take on one of only two possible values, the communication system is referred to as binary. If one of M > 2 possible values 

is available, it is referred to as M-ary. For long-distance transmission these 

digital baseband signals from the source may modulate a carrier before transmission. The result is referred to as amplitude-shift keying (ASK), phaseshift keying (PSK), or frequency-shift keying (FSK) if it is amplitude, phase, 

or frequency, respectively, which is varied in accordance with the baseband 

signal. More complex digital modulation schemes are also sometimes employed, 

but we will limit our attention to only these 123,241. 

A digital system is referred to as coherent if a local reference is available for demodulation which is in phase with the transmitted carrier (with 

fixed phase shifts due to transmission delays accounted for). Otherwise, it 

is referred to as noncoherent. Likewise, if a periodic signal is available at 

the receiver that is in synchronism with the transmitted sequence of digital 

signals (referred to as a clock), the system is referred to as synchronous; if 

a signaling technique is employed where such a clock is unnecessary the system 

is called asynchronous. A teletypewriter system is an example of an asynchronous system. 

The primary measure of system performance for digital data communication 

systems is the probability of error, pe. In this Appendix expressions for pe 

for various types of digital communication systems will be given. We are, of 

course, interested in receiver structures which give minimum pe for given background conditions. Synchronous detection in a white Gaussian-noise background 

requires a correlation or a matched filter detector to give minimum pe for 

fixed signal and noise conditions. 

E. 2. Baseband Transmission 

Consider the binary digital-data communication system illustrated in 

Figure E.2 where the transmitted signal consists of a sequence of constantamplitude pulses of either A or -A units in amplitude and r seconds in duration. A typical transmitted sequence is shown in Figure E.2(b). We may think 

of a positive pulse as representing a logic 1 and a negative pulse as representing a logic 0 from the data source. Each r-second pulse is called a bit 

for a binary digit. 

The channel is idealized as simply adding white Gaussian noise with 

double-sided power spectral density 1/2no to the signal. A typical received 
signal is shown in Figure E.Z(c). I!: is assumed that the starting and ending 

times of each pulse are known by the receiver. The problem of acquiring this 

information is referred to as synchronization. 

The function of the receiver is to decide whether the transmitted signal 

was A or -A during each bit period. A straightforward way of accomplishing 

this is to pass the signal plus noise through a lowpass predetection filter, 

sample its output at some time within each T-second interval, and determine 

the sign of the sample. If the sample is greater than zero, the decision is 

made that -i A was transmitted. If the sample is less than zero, the decision 

is that -A was transmitted. Since the starting and ending times of the pulses 

are known, a better procedure is to compare the area of the received signalplus-noise waveform (data) with zero at the end of each signaling interval by 

integrating the received data over the T-second signaling interval. Of course, 

a noise component is present at the output of the integrator, but, since the 

input noise has zero mean, it takes on positive and negative values with equal 

probability. Thus, the output noise component has mean zero. The proposed receiver structure and a typical waveform at the output of the integrator are 

shown in Figure E.3. For obvious reasons, this receiver is called an integrateand-dump detector. 

n Ct) : PSD = $no A 

Transmitter a 'L' Y\'-/ 

(+ A, - A) Receiver 

Figure E.2. System Xodel and Waveforms for Synchronous 

Baseband Digital Data Transmiss ion. (a) Baseband Digital 

Data Cornmunicat ion ,System. (b) Typical Trensnit ted Sequence. 

(c) Received Sequence Plus Noise. 
-fy=-l-Qq:+ Threshold > 0 Choose + 

I Signal plus noise 

7' 

Figure E.3. Receiver Structure and Integrator Output. 

(a) Integrate-and-dump Receiver. (b) Output From Integrateand-dump Detector. . - 

The integrate-and-dump receiver can be shown to be o?tinum for a baseband 

system. The analysis for the probability of error pe will not be given and 

can be found in a number of different texts. The baseband system is called 

unipolar if the two voltage levels are 0 and + A and bipolar if the two voltage levels are +A and -A. The probability of error is given in Table E.1, 

where S represents the signal power, the power density spectrum of the noise, 

N the total noise power, and r the pulse width; and B the bandwidth of the communication system. 

E.3. Digital Modulation Systems 

The use of direct digital transmissions is limited to low frequencies by 

the switching speeds available. In communication systems for various reasons, 

the digital signals are modulated on carrier signal before transmission. The 

three basic forms of digital modulation, are known as amplitude-shift keying 

(ASK) , frequency-shift keying CFSK) , and phase-shift keying (PSK) . The modulation methods will briefly be examined by their probability-of-error advantage. Although digital systems using more than two signaling states are sometimes used, most systems used are binary and the emphasis is on binary systems. 

E.3.1 Amplitude-Shift Keying (ASK) 

In amplitude-shift keying, the amplitude of a high-frequency carrier signal is switched between two or more values in response to the digital input. 
For the binary case, the usual choice is on-off keying (sometimes abbreviated 

as OOK). The resultant amplitude-modulated waveform consists of pulses, called 

marks, representing binary 1, and spaces representing binary 0. An ASK waveform is shown in Figure E.4 for a given code. The minimum bandwidth for an ASK 

signal can be shown to be doubled over a baseband system [23]. 

The ASK waveform for one pulse (i.e., a binary 1) can be written as: 

O<t<T C f (t) = t sin " ' otherwise 

The function of the receiver is to determine if a pulse was transmitted 

or not. The two most common methods of detection are coherent and envelope 

detection. The probability of error is less with coherent than envelope detection, however the latter is much cheaper to implement. The probability of error for coherent detsction is 

P e = 112 erfc &E 

and envelope detection is [23,24,25] 

E. 3.2. Frequency-Shif t Keying (FSK) 

In frequency keying, the instantaneous frequency of the carrier signal is 

switched between two (or more) values in response to the digital code. Figure 

E. 4 shows an idealized FSK signal. This suggests that one can consiaer the 

FSK waveform as composed of two ASK waveforms of diffzring carrier frequencies. 

Thus to convey either of the binary symbols, we have a choice of the two waveforms : 

0 

0 < t<T 

fl(t) = 

elsewhere 

O<t<T 0 

f2(t) = elsewhere 
The receiver must determine in this case which of the two frequencies was 

transmitted. Coherent detection may he used or two bandpass filters each 

tuned to the two separate frequencies and a comparator which samples the output 

of both filters determines if a "0" or "1" was transmitted. The coherent detector produces a probability of error 

'e = 112 erfc fi 

while the two filter detectors give 

The coherent detector gives a better probability of error, however the two filter method is much cheaper to implement 123,24,25]. 

E.3.3. Phase Shift Keying 

In phase shift keying, the phase of the carrier signal is switched between 

two (or more) values in response to the digital code. For the binary case a 

180' phase shift is a convenient choice becuase it simplifies the modulator design and hence is often used. The PSK signal can be written as 

fl(t) = A sinw t 

C 

f2(t) = -A sinw t 

C 

where fl(t) corresponds to logic "1" and f (t) corresponds to logic "0" or 2 vice versa. An example of PSK signal is glven in Figure E. 4. 

The only type of detector for a PSK signal must of necessity be a coherent 

type, since the information is contained in the phase of the waveform. The 

probability of error is 

'e = 112 erfc fi 

The complete tabulation of each pe for the different systems is given in Table 

E.1. There are other more sophisticated modulation systems with improved performance, but they are seldom seen in the mine environment and will not be 

examined. 
PHASE 

L(OWATrn 

Figure E.4. Modulation Techniques 
PROBABILITY OF ElXROR DEMODULATION TECHNIQUE 

Baseband 

Unipolar-Nonoptimum 

- 

pe = + erfc - = + erfcJS 2rl 2N 

ASK (OOK) 

- 

pe = erfc = + erfc~s 871 , N 

FSK 

Bipolar-Nonoptimum 

Unipolar-Optimum 

Bipolar-Optimum 

Coherent 

Noncoherent 

Coherent 

Noncoherent 

PSK 

- 

pe = 4 erfc JA 

2q 

=+erfc~z 

N 

Notation 

N = qB, q = power density spectrum, 8 = Bandwidth, N = Noise power 

T = signal interval, A = Level of signal 

Table 6.1. Probability of Error For Various Communication Systems 
y

code. A shortened cyclic code has the same error detecting capability as the 

code from which it is derived 128-311. The CRC-CCITT, CRC-16 and CRC-12 are 

in most cases shortened cyclic codes. Furthermore, they may be classified as 

shortened BCH cyclic codes in most cases. 

F.2. Notation 

It is assumed that anyone reading this is familiar with associating 

messages with coefficients of polynomials, and modulo two arithmetic involved 

in manipulating these polynomials. Otherwise, the following references are 

helpful [28-311. It is helpful to define certain symbols and polynomials. 

(These definitions are consistent with the literature.) 

k = number of binary digits (bits) in the message before 

encoding (sometimes called the number of information 

symbols). 

n = number of binary digits (bits) in the encoded message. 

n-k = number of check digits (bits) . 

b = length of a burst of errors. 
M(x) = Message polynomial (pf degree k-1) . 

G(x) = Generator polynomial (of degree n-k) . 

R(x) = Remainder on dividing xn-%(-XI hy G(-x) . R(x) is of 

degree less than n-k. 

T(.x) = Transmitted or encoded polynomial T(x) = xn-%(x) + 

R(x1 = G(x) Q(x). 

E(x) = Error polynomial (represents errors in transmission.) 

H(x) = Received encoded message polynomial H(x) = T(x) + E(x). 

[F(x) I = Number of non-zero coefficients in F(x) . 

F. 3. Princfples of Error Detection 

An encoded received message containing errors can be represented by 

where T(x) is the transmitted message and E(x1 is a polynomial which has a 

non-zero term (coefficient) in each erroneous position. Because the addition 

is modulo two, H(x) = T(x) + E(x) is the true encoded message with the 

erroneous positions changed. 

If the received message H(x) is not divisible by G(x), then clearly an 

error has occurred and retransmission is requested. If, on the other hand, 

H(x) is divisible by G(x), then H(x) is a code polynomial and we must accept 

it as the one which was transmitted, even though errors may have occurred. 

Since T(x) = G(x) Q(x), H(x) is divisible by G(X) if an only if E(x) is not 

evenly divisible by G(x). To insure an effective check, the generator polynomial G(x) must be chosen such that error patterns E(x) which occur the most 

often will not be divisible by G(x). For detection of random errors this is 

most effectively done by insuring all error patterns E(x) with d-1 or less 

non-zero coefficients are not divisible by G(x). (This is the same as insuring all pairs of encoded polynomials differ in d or more coefficients or 

are of distance d apart). 

F.4. Error Calculations 

BCH and other cyclic and shortened cyclic codes can be designed to 

detect all random error patterns with the number of errors in each pattern 

being 1,2, etc., up to a selectable numb,er of errors d-1. Therefore, an 

error pattern E(x) will only be undetected if d or more errors occur (i.e., 

number of non-zero coefficients is d or more in E(x)). It is assumed that 

synchronism is maintained between the sending and receiving units in all 

calculations of error probabilities. 

To determine the probability of undetected errors in a random channel 

(sometimes called a binary symmetric channel) it is necessary to define 

some terms. Let p = probability of a transmitted bit is received in error, 
P(undetected error)=P( I 

(x) I d,d  2,. . .n) = 1 In\ pm(lp)nm 

m-d \m, 

m even 

The BCH Codes are determined by their generator polynomial which are 

1 products of primitive polynomials that have a , a2,. . . ,ad-I as roots [28-311. 

Fortunately, tables exist for finding these polynomials. The generator polynomial for a number of protocols and various cases has been tabulated. Some 

protocols also call the BCH code a cyclic redundance code (CRC). The maximum 

length of the BCH Codes is znmk-l - 1,bits. If this length is exceeded, then 

the number of random errors detected will be less than the given values in 

Table F.1. The special case of parity checking can be represented by a 

special generator polynomial G(x) = x + 1. This error detection scheme is 

rather poor because not much data protection is given. Table F.1 also includes this case. 

F.5. Burst Error Detection 

A burst is defined as an error pattern in which the errors tend to occur 

in groups. The burst is defined by a length b in which the first and last 

bits are in error and the b-2 bits between the end bits may or may not be in 

error. The burst errors usually come from large transients such as switching, 

lightning etc. Any burst of length h - < n - k is detected with a generator 
polynomial of degree n - k. The fraction of bursts of length b > n - k that 

are undetected is (28-311. 

and 

2 Therefore, if G(x) = x16 + x15 + x + 1 then all bursts of length 16 or less 

are detected. This information is also given in Table F.1. 
Generator Polynomial 

Error D

Random Errors 

Detected 

Table F.1. 
-87- 

APPENDIX G 

DATA COMMUNICATION PROTOCOLS 

G.1. Introduction 

The great majority of contemporary data communications applications involve electronic machines as the sources and/or sinks for information being 

transmitted. The industrial and mine conununication and/or control applications 

are also becoming very sophisticated. Therefore, it is necessary that standard procedures and conventions govern the following: 

1. Message blocking and format organization 

2. Acknowledgment signalling and error control 

3. Terminal access to line 

4. Character synchronization between sources and sinks 

These procedures are called throughout the industry as communication line formats, conventions, line protocols, or line disciplines. Regardless of the descriptive term used, their objective is to provide for the systematic, 

unambiguous, orderly, reliable and generally automatic use of the communication 

links [36,371. 

The two most common new generation protocols are Digital Equipment Corporation's Digital Data Communications Message Protocol (DDCMP) and International Business Machine's Synchronous Data Link Control (SDLC). These two will 

be covered in detail because most new equipment uses one of these. 

G.2. Digital Data Communications Message Protocol--DDCMP 

DDCMP is designed to operate over clocked (synchronized) full- or halfduplex channels, switched or direct links, point-to-point or multi-point networks, and serial or parallel transmission facilities. Further, it will 

accommodate both synchronous and serial start-stop (asynchronous) modes. DDCMP 

is capable of controlling message transfers over standard existing hardware and 

can be implemented on many operating systems 1371. 

The format of a DDCMP transmission block is shown in Figure G.1. The only 

control character used in DDCMP is the first character in a message. It is 

used to distinguish between data, control, and bootstrap messages. SOH, ENQ 

and DLE are used, respectively. 

The header is required. It contains the count of 8-bit quantities (bytes) 

in the information field, some control flags, a response field for positive 

acknowledgment of received messages, a message sequence number (modulo 256), 

and an address. The latter field is mainly used for addressing tributary 

stations in multipoint configurations. The header is verified by having its 
ates in the full-duplex mode, the line does not have to be turned around. The 

NAK is simply added to the sequence of inessages for the transmitter. 

\?hen a sequence error occurs in DDC3IP, the receiving station does not respond to the message. The transmitting station detects from the response 

field of the messages it receives (or via timeout) that the receiving station 

is still looking for a certain message and sends it again. For example, if 

the next message the receiver expects to see is 5 and it receives 6, it will 

not change the response field of its data message which contains a 4. This 

says "I accept all sessages up through message 4 and I'm still looking for 

message 5." 

DDCMP uses the ASCII control characters SOH (Start of Header) ENQ 

(Enquiry), and DLE (Data Link Escape) to distinguish between types of messages. 

The remainder of the message (including the header) is transparent. This means 

the data in the information field is encoded by some higher-level protocol 

according to its design standard for information interchange. 

DDW achieves transparency by use of a count field in the header. The 

header is of fixed length. The count in the header determines the length of 

the information field which can be up to 16,383 bytes. To validate the header 

and count field, it is followed by a 16-bit CBC-16 field; all header characters are included in the CXC calculation. Once validated, the count is used 
STATION A 

(SENDING DATA) 

-89- 

STATION B 

(RECEIVING DATA) 

@ SENDS A STRT (START) b@ RECEIVES STRT 

MESSAGE WHICH MESSAGE 

MEANS: "I WANT TO 

ESTABLISH A CON- @ SENDS A STACK 

NECTIONWITH YOU (START ACKNOWI,- 

AND THE SEQUENCE EDGE) MESSAGE 

NUMBER OF MY FIRST ICH MEANS: "OK 

MESSAGE WILL BE 1." WITH ME; HERE IS THE 

FIRST SEQUENCE NUMBER (5) I WILL USE IN 

@ RECEIVES A STACK. e SENDING DATA MESSAGES TO YOU." @ SENDS DATA MESSAGES WITH A RE- @ RECEIVES DATA MESSPONSE FIELD SET TO SAGE 1 AND CHECKS 

4 AND THE SEQUENCE IT FOR SEQUENCE AND 

FIELD SET TO 1, WHICH CRC ERRORS. IF THERE 

MEANS: "I AM LOOK- IS A SEQUENCE 

ING FOR YOUR MES- ERROR, GO TO 12. IF 

SAGE 5 AND HERE IS THERE IS NO ERROR, 

MY MESSAGE 1." GO TO 9. 

OTHER MESSAGES 

MAY BE SENT AT THIS a A CRC ERROR WAS 

TIME (.I.E., MESSAGES DETECTED. COMPUTER 

2, 3, ETC.) WITHOUT B SENDS A NAK MESWAITING FOR A RE- SAGE WITH THE RESPONSE. SPONSE FIELD SET TO 

o , WHICH MEANS : I'ALL 

MESSAGES UP TO 0 

@ COMPUTER A RE- (MODULO 256) HAVE 

CEIVES NAK, RETRANS- BEEN ACCEPTED AND 

MITS MESSAGE 1 AND MESSAGE 1 IS IN 

ANY OTHER MESSAGES ERROR. 

SENT SINCE (I.E., 2, 3, 

ETC.) . \ 9 SENDS ACK RESPONSE 

OF SEPARATE 1 EITHER ACK IN MES- A

SAGE OR IN THE RE- 

@ RECEIVES ACK AND m SPONSE FIELD OF A 

RELEASES MESSAGE 1. DATA MESSAGE. 

@ SEQUENCE ERROR. 

SEND ACK RESPONSE 

CONTINUE SENDING -*?OF 0 IN THE RESPONSE 

MESSAGES. FIELD OF THE NEXT 

DATA MESSAGE TO BE 

/ SENT (IF THERE IS @ TIMES OUT BECAUSE NONE, THERE IS NO 

OF LACK OF RESPONSE RESPONSE). THIS 

FOR MESSAGE 1. RE- MEANS : tl~~~~FOR 

TRANSMITS MESSAGE 1. MESSAGE 1." 

Figure G.2--DDCMP Simple Handshaking Procedure (One ~irection) 
to receive the data and locate the second CRC-16 which is calculated on the 

data field. 

DDCMP uses either full-duplex or half-duplex circuits to their fullest 

extent. Both modes of operation are defined in a part of the protocol. In 

the full-duplex mode, it operates like two dependent one-way channels, each 

containing its own data-stream. The only dependency are the acknowledgments 

which must be sent in the data stream in the opposite direction. To reduce 

the response overhead, separate ACK messages are unnecessary. They are simply 

placed in the response field of the next message for the opposite direction. 

If several messages are received correctly before its transmitter is able to 

send a message (because the previous message was a long one), all of them can 

be acknowledged by one response. Only when a transmission error, occurs or if 

traffic in the opposite direction is light (no data message to send) is it 

necessary to send a special NAK or ACK message, respectively. 

In summary, DDCMP line utilization features include: 

1. Th-e ability to run on full or half-duplex transmission facilities. 

2. The ability to run on many existing hardware interfaces. 

3. The ability to support point-to-point and multipoint lines. 

4. Transmission mode independence Csynchronous, asynchronous, or parallel). 

5. No separate ACK1s when traffic is heavy. 

6. Multiple acknowledgements per ACK (up to 255 messages in one acknowledgment). 

DDCMP achieves synchronization through the use of the two ASCII SYN characters preceding the SOH, ENQ, or DLE. It is not necessary to synchronize between messages as long as no gaps exist. This feature helps reduce the control 

overhead because each SYN takes 8-bit times. Character synchronization is 

unnecessary when DDCMP is used for serial asynchronous and parallel channels. 

DDCMP can be used for serial synchronous, serial asynchronous, and parallel facilities. This is total facility transparency. 

G.3. Synchronous Data Link Control--SDLC 

IBM1s SDLC (Synchronous Data Link Control) was announced in 1973 and operational on products in 1974. Unlike DDCMP, it is bit-oriented rather than 

character-oriented. It is designed for full and half-duplex operation [36]. 

The format of SDLC is shown in Figure G.3. The only control character 

used in SDLC is the flag character which has the bit pattern 01111110. There 

is a fixed length 24-bit header, a variable length information field, and a 

fixed-length 24-bit trailer. In addition, the transparency technique 
(explained below) can increase the size of any of these fields except for the 

two 8-bit flags that frame the message. 

SYMBOL DENOTES 

START & END OF FRAME 

I 

Figure G .3--SDLC Message Format . -- - ---. -. - 

Like DDCN?, SDLC may operate on full-duplex facilities and uses an efficient procedure for data exchange. The example in Figure G.4 presents only a 

one-directional transfer of data in a full-duplex operation. Equivalent operation in the other direction also occurs because it is symmetrical. 

FLAG 

8BITS 

SDLC uses CRC-CCITT to detect transmission errors. It handles CRC with 

an inversion technique that differs from methods used by character-oriented 

protocols. This technique improves the range of undetected errors in the domain of possibilities. Like DDCMP, SDLC has a response field (3 bits in the 

control field) and separate ACK and NAK messages. Unlike DDW, SDLC does not 

NAK transmission errors. The way it handles them is best explained by example: 

Station B receives messages 2, 3 and 4 from Station A and message 4 is bad. 

Station B in its next data message for Station A, responds with message 4 

which says: "I have received and accepted messages 2 and 3 and I am still 

looking for message 4". Station A, knowing it has sent message 4, must wait a 

period of time to see if it will be acknowledged (timeout) before it sends the 

message again. 

Unlike DDW, SDLC responds to sequence errors with a NAK message (a less 

frequent occurrence than transmission errors). The response is sinilar to 

DDCXP's handling of transmission errors in that the NAK acknowledges the messages received in correct sequence condition, i.e., "received 4, 5 correctly 

and looking for 6" when it receives 7 instead. 

ADDRESS 

8BITS 

SDLC is not concerned with information exchange codes. The only control 

character is the flag and it has a fixed pattern (01111110) regardless of the 

exchange code used by the systen. Like DDCXP, a higher-level protocol defines 

the information exchange code used to transfer meaningful data. 

SDLC is a bit-oriented protocol and only has to be sure that a flag character bit pattern anywhere between frames does not arrive at the receiver. 

CONTROL 

8BITS 

The technique used by SDLC for achieving transparency may be referred to 

as "bit stuffing." SDLC uses a character at each end of the message called a 

"flag." The flag's bit sequence is 01111110. The only control characters are 

the flag characters. To ensure that a flag character does not appear in the 

data portion of the message, a zero (-0) bit is inserted whenever five one (1) 

INFORMATION 

ANY NUMBER 

OF BITS 

FW-W CHECK 

SEQUENCE 

16 BITS 

FLAG 

8 BITS 

I 

i 
STATION B 

(SENDING DATA) 

STATION B 

(RECEIVING DATA) 

SENDS AN SNRM (SET 

NORMAL RESPONSE a RECEIVES SNRIf. MODE) MESSAGE 

WHI CH MEANS : "YOU 

MAY NOW START UP. a SENDS AN NSA (NON- MESSAGES TO BE 

SEQUENCED ACK 

WHICH MEANS: 

SENT AND RECEIVED 

ARE NLPBERED 

RECEIVED YOUR STARTING AT 0. I' 

SNRM. " RECEIVES NSA 

RECEIVES DATA MESSAGE 0 AND CHECKS 

THE MESSAGE FOR @ SENDS A MESSAGE SEQUENCE AND CRC 

WITH RESPONSE FIELD ERRORS. IF THERE IS A 

SET TO 0 AND THE CRC ERROR, GO TO 12. 

SEQUENCE FIELD ST IF THERE IS NO 

TO 0 WHICH MEANS : ERROR, GO TO 9. 

"I 'M LOOKING FOR A SEQUENCE ERROR 

YOUR MESSAGE 0, WAS DETECTED. COMAND HERE IS MY PUTER B SENDS AN 

MESSAGE 0." REJ (REJECT) MESOTHER MESSAGES SAGE WITH THE RECAN BE SENT AT THIS SPONSE FIELD SET TO 

TIME (I .E. , MESSAGE 0. THIS MEANS: "I 

1, 2, ETC.). WITHOUT HAVE ACCEPTED ALL 8 

WAITING FOR A RE- MESSAGES PRIOR TO 

SPONS E . 0 (MODULO 8) AND I 

AM STILL EXPECTING 

MESSAGE 0." @ COMPUTER A RE SENDS ACK RESPONSE 

CEIVES REJ, RETRANS- OF I EITHER IN A 

MITS MESSAGE 0, AND SEPARATE CONTROL 

ANY OTHER MESSAGES MESSAGE OR IN THE 

SENT SINCE (I.E., 1, 2, RESPONSE FIELD OF 

ETC.). THE NEXT DATA 

MESSAGE TO BE 

SENT. THIS MEANS: 

"I'M LOOKING FOR @ RECEIVES ACK MESSAGE 1." GO TO 

RELEASES MESSAGE 0. 6 EXPECTING MESSAGE I. 

CRC ERROR. SEVD ACK @ CONTINUES SENDING - RESPONSE OF 0 IN THE 

MESSAGES. RESPONSE FIELD OF 

THE NEXT DATA MES- 0 TIMES OUT BECAUSE SAGE TO BE SENT (IF 

OF LACK OF RE- THERE IS NONE, 

SPONSE FOR MESSAGE THERE IS NO RE1. RETRANSMITS MES- SPONSE) . THIS NEANS: 

SAGE 1 AND OTHER "I'M LOOKING FOR 

MESSAGES. MESSAGE O.I1 

Figure G.4--SDLC Simple Handshaking Procedure (One Direction) 
bits appear in a row. Thus, a OllllllO bit pattern meant to be data would 

appear in the data stream as Ollll1010, or 9 bits. The receiver also counts 

bits, and if it detects five one's in a row followed by a zero bit, it removes 

the zero bit. If it was a one bit, it is a legitimate flag and the end of 

message has been received. 

SDLC uses transmission facilities with similar efficiency to DDCMP because of the following features: 

1. The ability to transmit on full or half-duplex facilities. 

2. Low control character overhead 'flag, header, and check bits 

total 6 characters). DDCMP uses 10 characters 1281. 

3. No "character stuffing"; SDLC uses "bit stuffing". In contrast, 

DDCMP's transparency overhead is fixed because it used a count 

method. 

4. No separate ACK messages are necessary. 

5. Multiple acknowledgments per ACK Cup to 7 messages). 

6. The ability to support point-to-point and multipoint lines. 

SDLC synchronizes on the flag characters between messages. No SYN characters are required. 

Because of the "bit stuffing" requirement of SDLC, it cannot be used for 

serial asynchronous or parallel facilities. Asynchronous characters are fixed 

length and "bit stuffing" would destroy them. On parallel connections, a 

separate wire(-s) would be needed for the stuff bit (s) . SDLC is designed for 

efficient use of high speed serial synchronous full-duplex facilities. 

G.4. Other Protocols 

A number of national and international standards organizations have been 

cooperating over the past years to produce standard data communication protocols. These organizations include the American National Standards Institute 

CANS11 and the International Standards Organization CISO). 

The cooperative efforts of these organizations have resulted in the development of a variety of standard data communications control procedures and 

codes. Typical of these developments are the Advanced Data Communication Control Procedures (ADCCP) developed by ANSI, and the High-Level Data Link Controls (-HDLC) developed by ISO. These protocols differ slightly from SDLC. 

A number of manufactures of communication equipment for control and monitoring have developed relatively specialized protocols for their own equipment. 

Rarely can more than one manufactures equipment be tied into the communication 

system. Conspec Protocol supports asynchronous transmission, uses a parity bit 

for every 8 bits of data, and sends two characters during each transmission. 

This protocol is obviously not as comprehensive as DDCMP or SDLC. 
The result of a transmission verification algorithm accumulated over a 

transmission block, and normally appended at the end. 

Carrier 

A continuous frequency capable of being modulated or impressed with a 

signal. 

Channel 

Part of a cormnunications system that connects a message source to a 

message sink. 

Coaxial cable 

Cable consisting of a conductor completely surrounding another conductor 

and separated from it by a solid dielectric or air with spacing insulators. 

Cyclic Redundancy Check CCRC) 

An error detection scheme in which the check character is generated by 

taking the remainder after dividing all the serialized bits in a block of data 

by a predetermined polynomial. 
Data Link 

An assembly of terminal installations and interconnecting circuits operating according to a particular method that permits information to be exchanged 

between terminal installations. 

Field 

An area under the influence of an electromagnetic or electrostatic charge. 

Frequency Shift Keying (FSK) 

A modulation scheme in which the instantaneous frequency may take on only 

two possible values. 

Modulation 

The process of superimposing a signal on a carrier wave so that it may be 

transmitted. 

Noise 

Any undesirab-le disturbances in a communication system. 

Parallel transmission 

Method of data transfer in which all bits of a character or byte are 

transmitted simultaneously either over separate communication lines or on different carrier frequencies on th.e same communication line. 

Phase modulation 

A method of transmission whereby the angle of phase of the carrier wave 

is varied in accordance with the signal. 

Protocol 

A formal set of conventions governing the format and relative timing of 

message exchange between two comunicating processes. 

Reverse channel 

A channel used for transmission of supervisory or error-control signals. 

The direction of flow of these signals is in the direction opposite to that in 

which. information is being transmitted. 

Serial transmission 

A method of transmission in which each bit of information is sent sequentially on a single channel rather than simultaneously. 

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