Contract J0123030 

University of Michigan 

Cooley Electronics Laboratory 

BUREAU OF MINES 

UNITED STATES DEPARTMENT OF THE INTERIOR 
The views and conclusions contained in this document 

are those of the authors and should not be interpreted as 

necessanly representing the official policies or recommendations of tne lntenor Department's Bureau of 

Mines or of the U.S. Government. 
PAGE 

~ntl..ndsmtW 

I 

%-- 

December 198 3 I Field Test of a Post-Disaster Communication System I 

I 

7. krChorU 

M. P. Ristenbatt, E. K. Holland-Moriltz, K. Metzger 

r--W.m-dmd?mm 

Cooley Electronics Laboratory 

The University of Michigan 

Ann Arbor, Michigan 48109 

=3anursrl~)(.rrwad~. 

U. S. Bureau of Mines 

Pittsburgh Mining & Safety Research Center 

Pittsburgh, Pennsylavania 15213 

1L-w~)Ir. 

* 50123030 

(m 7 

lSnmdR...rlLCIWh 

Final 

a-(Us*aa~-~ A new post-disaster signalling system for the Bureau of 

Mines, using through-the-earth EM propagation, was successfully fieldtested in the Lake Lynn test mine. Suc:cessful detection at an estimated 

input SNR of -30dB was accomplished using a coherent integration time of 

4.46 minutes. 

The transmitted waveform consisted of a repeating pseudonoise maximal 

length sequence phase modulating a 1950Hz carrier. The receiver used a 

I sliding-window -matched filter, optimal for signal known exactly except for 

phase and starting time, implemented vi.a a special purpose processor. I 

Parallel processing permitspadaptive ~i.~nallin~, where the detection (integration) time is determined by the ac!tual input SNR encountered. 

I This field test demonstrates a tra.pped miner detection system that is 

about 36dB more sensitive than a current pulsed system. This increased 

sensitivity permits some desired combination of increased range, shorter 

detection time, and reduced transmitter power or antenna-size. The combination of pulse-compression-type-signal, coherent integration, parallelprocessing, and output peak-detection algorithms permit implemention of an 

ultra-reliable detection system. The same combination is compatible with 

digital Pulse Position Modulation and Eimitter position location strategies. 

Mine Communications Eme:rgency Communications 

Trapped Miner Communication Dig.ital Communications 

Adaptive Communications Through- the-Ear th-Propagation 

Spread Spectrum High TW Signaling 

Feedback Communications Matched Filtering 

Coherent Detection Maximal Binary Sequences 

Signal Processing 

c -m CkY/chum L 

, ir Arsrrrarrw 

t 

c3me-lY) b 

--o;r4tm 

U.~uaamblbCrd 

Unclassified 

jas-uWQlr~J'-em) 

Unclassif~ed 

1 

ZLPkd51 

tLmw 
FOREWORD 

This report was prepared by The University of Michigan, 

Cooley Electronics Laboratory, Ann Arbor, Michigan under USBM 

Contract number 58123030. The contract was initiated under the 

Coal Mine Health and Safety Research Program. It was 

administered under the technical direction of the Pittsburgh 

Mining and Safety Research Center with Steven Shope as Technical 

Project Officer. Doyne W. Teets w'as the contract administrator 

for the Bureau of Mines. This report is a summary of the work 

recently completed as a part of this contract during the period 

June 1982 to September 1983. The draft report was submitted by 

the authors in December 1983. 

Steven Shope was responsil~le for a11 the physical 

arrangements for the data taking, and supervised the analog 

recording at the surface. He operated the transmitter within the 

mine, and provided FFT-type analysis results of both transmission 

and surface-received signals. 

The authors wish to acknowledge the technical discussions 

with T. G. Birdsall of CEL's Signal Processing Group, and 

acknowledge the use of his (ONR sponsored) signal processing 

equipment and expertise. The availability of these items, at no 

cost to the Bureau of Mines, made it possible to accomplish 

significant tests with a minimum budget. 
- TABLE OF CONTENTS 

Page 

DOCUMENTATION PAGE 

FORWARD 

SECTION 

1. Introduction 

2. Background 

3. Equipment Preparation 

3.1. PN Encoder 

3.2. Signal Processing 

4. Test Configuration 

4.1. Physical Site and Arrangement 

4.2. Field-Test Recording and Monitoring 

4.3. Test Series 

5. Specific Signal Processing Methods Used 

5.1. Off -Line MTXC Processing Arrangement 

5.2. Special Processing for Displays 

5.3. Special Clipper Signal Processing 

6. Input SNR Aspects 

6.1. Noise Types 

6.2. Input SNR Estimation 

7. Test Results 

8. Conclusion and Recommendat ions 

APPENDICES 

A. A Sample of the Printed Data Output 

of the PN Processor 

B. Comparison of the Pulsed System to 

the PN Method 

C. input SNR Estimates 

REFERENCES 
FIELD TEST OF A POST DISASTEF! COMMUNICATION SYSTEM 

M. P. Ristenbatt, E. K. Holland-Moritz and K. Metzger 

1. Introduction - --- 

The objective of this effort was to document the field 

performance of a post disaster mine signalling system that 

recently has been demonstrated in the laboratory (Ref. 1, 2). 

The three major objectives of our effort were: 1) prepare 

transmitters by designing and adding a new signal generator to an 

existing Bureau of Mines transmitter; 2) cooperate with the 

Bureau of Mines in obtaining recorded data during field tests; 

and 3) process and analyze the tape recorded data from the field 

test. 

2. Background - --- 

The Burem of Mines currently has a signalling system which 

is intended to permit a trapped miner to transmit a periodic 

signal to the surface, or to an interior receiver, using 

through-the-ear th electro-magnet ic wave propagation. The 

transmitted signal is a low audio frequency pulse with a duration 

of 0.1 second which repeats every second. The frequency of the 

tone is selected t.o fall between the harmonics of 60 Hz, which 

are significant background noise contributors. The receiver 

picks up the signal on a loop antenna and translates it to 1600 

Hz. The signal is narrowband filtered and presented to the 

operator by earphones and a visual meter indicator. The output 

of the receiver consists of the noise from the (30 Hz-wide) 

carrier frequency band translated to 1000 Hz accompanied by 

possible l00@ Hz signal pulses. Detection in a noisy background 

depends on a number of successive signal pulses being high enough 

above the (1000 Hz) background noise so that the operator can 

recognize the 1 second spacing. In effect, the receiver consists 

of a "front-end' bandpass-amplifier , in combinat ion with the eer 

(or eye) plus brain. 

After a signal is detected, the operator uses horizontal 

(walking) displacement, and rotation of the loop to locate nulls 

in the vertical or horizontal magnetic field of the transmitter. 

This allows locating a point directly above the transmitter. 

In our predecessor laboratory demonstration work (Ref. 2) , we verified, using observer tests and theoretical models, that in 
Gaussian noise, this system requires an input signal-to-noise 

ratio (SNR) of about 6 dB for specified detection performance. 

A prior study (Ref. I), showed thzt signal detection at a 

much lower SNR would be possible in the Mine Disaster 

Communication system if; 1) the 1/1W second carrier pulse, 

repeated each second, were replaced with a continuous carrier 

phase modulated by a rzpeat ing binary pseudo-noise (PN) maximal 

sequence; and 2) the human operator with earphones were replaced 

by a digital matched filter for coherent integration of one or 

more periods of the binary PN signal. 

Important aspects of the new system are: 

1) coherent integration of the signal "known exactly except 

for phase and starting time," over time intervals from 

(say) 17 seconds up to (say) 17 minutes. 

the FN (maxima?-length sequence) signal implements a 

large time-bandwidth product signal, having an ideal 

autocorrelation. The trznsmitted PN signal is a 

constant amplitude phase-modulated signal. At the same 

time, the receiver output, after matched filtering, is a 

pulse occuring at one of 255 possible pulse positions. 

This provides: (a) a practical way for observing the 

actual noise background dur ing signalling , at the times 

between pulses. (b) the ability to receive up to 255 

different signals without interference in one 3@ Hz 

bandwidth based on different PN sequence starting times 

(assuming no oscillator drift). These could be alarms 

from other miners or signal codes used for communication 

during rescue operations. (c) a receiver which can be 

made more resistant to both pulse and CW interference 

than the present pulse receiver. (d) a continuous 

transmitted signal which can be measured by a simple 

receiver for locating nulls, after the signal phase and 

sequence starting time are determined by an alarm 

receiver . 

3) a parallel receiver processing is used which 

simultaneously tests for potential alarm signals after 

integration over 1, 4, 16 and 64 sequence periods. This 

allows rapid response to strong signals, without the 

risk of losing alarms under less favorable conditions. 

In the immediately preceding work (Ref. 2) , we demonstrated 

in the laboratory successful operation in white Gaussian noise at 

an input SNR of -34 d8.l This performance required coherent 

'~aussian noise is the worst case, since it is unpredictable 

except for its statistical properties. The actual mine noise 

includes periodic components and pulses which can be suppressed 

more easily than Gaussian noise. 
integration of 17 minutes. This maximum potential decrease of 

required input SNR by 40 dB, from the 6 dB of the present system 

to the potential -34 dB of the proposed system, permits some 

desirable combination of: 1) reducing the trsnsmitter current or 

antenna magnetic moment; 2) increasing the reliability and/or 

extending the system range; 3) using a moderate detection time 

(say 17 seconds), with automatic back-up for less Eworable 

conditions; and 4) increasing the accuracy or reliability of 

null location in noisy backgrounds. 

The PN maximal sequence is a particularly favorable choice 

of bits for efficient transmission because it has a constant 

envelope and an ideal pulse autocor relation function. The output 

of the PN matched filter for each 255 chip1 sequence is one 

signal pulsz and 254 other possible pulse positions where noise 

measurements can be made, or ather PN signals received. The 

constant envelope of the PN signal allows the output pulse power 

for 16 sequence repetitions to be (16x255)=36 dB highec thzn the 

peak pulse power for a TW=1 signal. 2 

Additional improvement in the performance of the PN system 

is possible if the noise is non-Gaussian. For example, if the 

noise consists of isolated pulses, only that portion of the pulse 

power which falls in the PN signal bandwidth appears zt the EN 

output. The same matched filter processing which converts the PN 

transmitted signal to a pulse output, tzkes the pulse power in 

the PN bandwidth and spreads it uniformly over the PN signal 

integration time. If the PN pulse detection system compares 

possible pulse positions with the background noise, it only 

measures an increased continuous background, without pulses, no 

matter how large the power in the isolated pulse. False alzrms 

due to isolated pulse noise can occur only if there are many 

large pulses in the PN integration period. When this is the 

case, the constant envelope of the PIN carrier can be exploited by 

using a limiter ahead of the PN matched filter to limit the peak 

received input signal amplitude to tlhe (observed in some fashion) 

amplitude of the PN signal plus background Gaussian noise. Tnis 

reduces all pulse noise to small pulses. The residual effect of 

many clipped pulses is both to reduce the signal peak and to 

increase the effective Gaussian noise background. Nevertheless, 

a limiter is very effective in a PN system especially when large 

or unpredictable noise pulses are present. 

A limiter is much less effective in a (TW=l) pulse system, 

because it cannot limit the noise without limiting the peak 

output pulse to the same level. with PN processing the output 

'A "chip" means one modular part of a total signal; the number 

of chips essentially means "number of independent samples in the 

signal," and hence, equals the time-bandwidth (TW) product of the 

signal ( 255 here) . A simple pulse si-gnal, as in the present system, has a nominal TW product of 1. 

2~n this application, the peak transmitter power is limited both 

by the miner's battery and by mine sa~fety restrictions. 

8 
signal pulse power is increased by a factor equal to the TW 

product after the limiting. 

It may not be obvious, but a very similar suppression of 

isolsted CW interference can occur in a well designed PN system. 

This type of interference can occur from harmonics of portable 

power generators operating slightly off 60 Hz frequency, 

harmonics generated by induction motors, and some types of 

starting and stopping transient.^. This exploits the PN-signal 

nulls at harmonics of 60 Hz (see Fig. 3 of Ref. 2). 

From the above, we see that the noise-like PN matched filter 

is much more efficient in suppressing both the pulse and the CW 

interference than the matched filters for either pulse or CW 

signals alone. It simultaneously combines the best features of 

both, and adds some features of its own. Unless the interference 

is extremely strong, this may be all the protection which is 

needed. As will be described, on the second mine test day, the 

pulse and pulse burst interference was very strong. In this 

case, s limiter ahead of the PN filt2r can reduce the noise at 

the system output as was demonstrated later in our off-line 

processing results. 

After an alarm signal is detected, the PN signal used here 

is an excellent signal for digital data transmission. Hence, 

post-detection communication may utilize a real time voice 

downlink with either a digital data or an off-line digital voice 

uplink. 

- 3. Equipment Preqaration --- - --- 

- 3.1. - PN Encoder 

As noted, the University of Michigan Post Cisaster 

Communications System uses a continuous carrier phase modulated 

by a digital pseudo-noise signal. 'The transmitter for the field 

tests was implemented by adding a new signal source to the 

existing Bureau of Mines pulsed VF transmitter (Circuit AlP2, 

General instrument Ref. 3). 

As specified by the contract, two of the transmitters 

available for use in the present system wer? modified to generate 

and continuously transmit a 1950 Hz carrier phase modulated by a 

255 chip-long maximal sequence, at a chip-ratz of 15.23 

chips/second.' There are 128 carrier cycles in the 65.64 ms-long 

l~he signal spectrum nulls could be centered exactly on the harmonics of 60 Hz by using a 15 Hz chip-rate and 32.5 cycles of carrier per demodulate. This would be compatible with the present 

receiver algorithms. However, using an integral number of cycles 

per demodulate may be slightly more efficient in rejecting low frequency noise and A/D offset errors. We chose to use 32 cycles per 

demodulate, but 32.5 is a viable opti-on. 
sinx 2 chip. The resulting signal power spectral density has a (- X 1 

envelope with a main lobe width of 3Q.46 Hz. It lies midway 

between the 32nd and 33rd harmonics of 6ti Hz and has broad nulls 

at those harmonics. 

The choice of a 255-long sequence, made in the original 

design effort (Ref. 1) , considered the background noise to be 

Gaussian and was based simply on the criteria that we desired a 

sufficient number of "noise-only" ssmple values between output 

peaks (when the signal is present), and the operztionsl 

considerat ion of minimum decision-delay time. It was assumed 

that RF-filtering would suppress any harmonics of 60 Hz adjacent 

to the carrier. Further experience convinced us thet the 

filtering should be done by the signal processor itself, so that 

some harmonics of 60 Hz may be present at the processor input. 

It may be advantageous, from the siqnal processor point of 

view, to consider a sequence-length of 63. It is not yet clear 

whether this change would decrease the processor gain against the 

harmonics, but will be studied in the future. 

Figure 1 shows the General Instrument circuit diagram (their 

Figure 2-2, p. 11) from their final report (Ref. 3), along with 

the PN encoder connection points. A carrier frequency of 1550 Hz 

was recommended by the Bureau of Mines, based on considerations 

of both propagation loss and noise behavior. The logic circuit 

diagram of the PN encoder that was designed is shown in Fig. 2. 

In the previous laboratory test (Ref. 2), our transmitted signal 

was a sinusoidal carrier which was digitally generated 2nd 

digitslly phase modulated. The Bureau of Mines transmitters 

supplied for the present contract requires a binary digital input 

and supplies a square wave carrier voltage to the transmitter 

loop antenna. After filtering by both antenna systems and the 

receiving filters, the received signal carrier is sinusoidal in 

both cases. An improvement, relative to the prior-effort signal 

generator, was made in the efficiently of the PN signal generated 

by the new boards. In our previous contract (Ref. 1) , the PN 

phase modulation of the carrier was 160'. Since there are an odd 

number of bits in a maximal PN sequence, this actually results in 

a signal with a reduced value at the carrier relative to the 

adjacent signal sidebands. In tlne time domain, the receiver 

output has an off set (minus-one) between the pulse-compressed 

auto-correlstion peaks. In the new board the phase modulation 

angle is 172. eO, which eliminates the minus-one off set between 

peaks, and produces a processed output signal with the ideal 

zero-r2ferenced autocorrelation function. This technique was 

developed by Prof. T. G. Birdsall znd Cr. Kurt Metzger in 

connection with their long-standing work in utilizing PN seauence 

signals for studying the properties of underwater sound 

propagation, and is thoroughly detailed in Ref. 4. 

Coherent integration, in the absence of strong carrier, 


requires clock stability at both transmitter and receiver that is 

commensurate with the maximum integration time. The basic 

criterion is that there be substantially less than 180' phase 

shift, at the carrier freuuency, over the total integration 1 

I time. 

If we upper bound the integration time as being in the order 

of 10 minutes, then the total number of "parts" involved is: 

(1950 w) x 600 sec. = 1.17 x lo6 total 

sec, 

If we choose to keep the phase-shift to about 10% of one 

part in the 10 minutes, then the required stability is: 

. 1 Req. Stability = -------- - .85 x lfl-7 I x l~-~ (2) 

1.17 x lo6 

Thus, a stability of one part in 10~ was selected for these test 

transmitters. 

This criterion led to our selecting the Vectron Labs (CO252B17V) crystal oscillator, which , ground to our custom 

frequency, cost $410.00, This cost %would decrease with volume 

purchases. 

The oscillator module and PN phase modulator ace mounted on 

a circuit board similar to the present transmitter boards as 

shown in Fig. 3 and in the photograph of Fig. 4. This unit could 

fit into the present VF transmitter boxes above the existing 

circuit board. However, since the power transistors on the VF 

transmitter board are a significant source of heat, we thought it 

desirable to mount this new board in a separate box, to isolate 

the crystal thermally. This may not be necessary in practice, 

but appeared to be a sensible precaution at this field-testing 

phase. - (One of our sgcondary objectives was to exploit the 

natural opportunity to determine the source of any limitations). 

This arrangement allowed us to distinguish, in the laboratory, 

between residuzl frequency drift due to temperature variations 

and frequency drift due to supply voltage variations. 

- 3.2. --- Signal Processing - 

The general concept for the signal processing was described 

in the report for the predecessor effort (Ref. 1) , and will only 

I The long term stability requirement for the transmitter can be 

relaxed, but only at the cost of increased receiver complexity. 

More complicated algorithms for underwater acoustic signals can 

detect doppler shifted PN sequences with unknown frequency shifts. 

The phase of the received signal as a function of time was in 

fact displayed as part of each of the present tests. However, no 

correction for the phase errors was included in the processing 

algorithm. 

13 


be synopsized here. The essential idea is to first demodulate 

the signal using the traditional complex demodulator, to generate 

4 inphase and 4 quadrature demodulates per sequence chip (a 

sequence bit is one "chip" of the entire integrated siqnal) . The 

dig it31 matched filter processes the inphase and the quadrature 

components separately in increments of 1/4 sequence chip. The 

matched filter outputs for successive periods are added 

coherently for various integrating times: 1 sequence-length 

(16.74 seconds) ; 4 sequence-lengths (66.954 seconds) ; 16 

sequence-lengths (4.46 minutes) ; and 64 sequence-lengths (17.854 

minutes). For each integration time, the 4 demodulates per chip 

are averaged by a sliding window. The inphase and quadrature 

components are then combined as the sum of squares. Processing 

the received signal simultaneously with different periods of 

coherent integration permits a receiver to "alarm" at the 

shortest time, after onset, commensurate with the received SNR. 

With the parameters chosen here, an alarm could occur t 17 

seconds after onset if the SNR is adequate, and at one of the 

longer times for lower received SNR1s. 

The signal processing software and hardware used to process 

and display the PN signals were small special purpose computers 

progrzmmed to demodulate the signal and then to operate as 

matched filters either for the sequence itself, or for a periodic 

signal consisting of 4, 16 or 64 repetitions of the PN sequence. 

The output of the digital matched filters is a "time compressed" 

pulse with a (two-sided) duration of twice the chip length 

occurring once every 16.54 seconds. 

The actual equipment used for processing and display was 

designed and constructed by the CEL Signal Processing Group 

mentioned before (T. G. Birdsall and K. Metzger) . The equipment 

was designed and developed for processing acoustic signals in 

Oceanography. It was reprogrammed to match our signal and 

display parameters. 

Figure 5a repeats (from Ref. 1) the general classical 

concept for matched filtering to a "signal known except for phase 

and starting time," for the particular signal parameters used 

here. In effect, the entire PN signal, over 1, 4, 16 or 64 

sequence epochs, is known except for the uncertain turn-on plus 

propagat ion-delzy time. The proper receiver, in terms of 

classical communication pr inciples, would be called "a complex 

demouul-ator for a large time-bandwidth-produc t signal." 

The signal is first band limited in a 1/10 octave filter. 

As indicated in Fig. 52, the signal is then sampled and A/D 

converted at four times the carrier frequency (4 x 1950 = 7800 

samples per second). An inphase demodulate, corresponding to 

multiplying by the c0s2a (1950) t, is obtained by forming the sum 

of the first samples of each cycle for 32 consecutive cycles, 

then forming the sum of the 3rd samples for the same 32 cycles, 

and subtracting the two sums. Similarly, a quadrature demodulate 
- A-D - 

Sample at 

Digital i + Matched Filter 1 

(a) Classical Concept of Complex Demodulator and Parallel Processing 

(to 255-Chip 

- 

Figure 5. Block Diagram of Signal Processii~g Arrangements 

I 4 

Square 

PN Sequence) , A 

Cos 2n (1950)t J 

-Sin ?IT (1950) 

, 8 

Parallel 

Processl for 

1,4,16,64 

PN Periods 

Digital 

, Matched Filter 

(to 255-Chip 

Q PN Sequence j 

Display 

Time 

Axis 

0 

0 , 0 

. 9 

I P 

, Square 

. 

, - 
is derived, corresponding to mu1.t iplying by- -sin2~(1350) t, by 

forming the sum of the 2nd samples for 32 cycles, then forming 

the sum of the 4th samples for 32 cycles, and subtracting the two 

sums. The inphase and quadrature demodulates are timeintzrleaved, and zppear sequentially at the output as 8 real 

numbers per chip. 

During the predecessor-effort laboratory demonstration, the 

complex demodulation processing and recording the demodulates on 

digital tape was done in a CEL computer. The matched-filter 

integrating was done of f-line on the University of Michigan timesharing MTS AMDAHL computer. For this present effort, we were 

able to exploit further equipment development by Messrs. 

Birdszll and Metzger. In particular, a new portable processor 

(called DM12aB) became available. The DM120B is a special 

processor (17" x 17 1/2" x8 1/2") designed and built at CEL, 

using standard mother-board construct ion, with a 21K byte memory. 

For off-line processing, the EM1200 forms the complex demodulates 

and performs the matched filter operations. 

As indicated in Fig. 5b, the DM1200 was used twice in the 

off-line processing. The MTDEMOD program was used first, to form 

demodulates which are stored on digital reel-to-reel tape. Then, 

using a tape playback for these demodulates, the EM1200 is used 

again with an MTXC program which cross-correlates the demodulates 

with the PN sequence reference. The output is an uncombined 4 

chips per bit, and is essentially the matched filter or 

correlated output for a single PN sequence epoch. This is again 

stored on a digital reel-to-reel tape. In effect, the 

correlation slides the reference-sequence along the data samples 

in increments of 1/4 chip resolution. This off-line processing 

permits detailed examination of the data. 

A final step at the time-shared U-M Computing Center first 

coherently combines the 4 chips per bit, 2nd then coherently 

averages the results over 1, 4, 16 and 64 FN sequence epochs. 

The filtered inphase and quadrature samples for each sliding 

window position are then squared and added for the final output. 

A print-out program permits tallying the entire sequence of 

output numbers, for each instant, which provides quantitative 

data for the peak values and the intervening noise-only values 

(see Appendix A) . 

Perhaps the most important role of the DM1208 was its use of 

another program for real-time processing (see Fig. 5c). This 

permitted us to process the field data in real time to find the 

operating threshold for the system. As noted in Fig. 5c, the 

real-time processing used the SCMP program in the DM1208 

processor, which produced a graphical display on a CRT which 1 i could be copied on a printer. 

'~t the time of signal processing, this program did not provide 

numerical data for the output SNR fo:r each trace. These values 

were later added to the other data printed under each trace. 

[I) 

c, 

- fi 

[I) 

[I) 

Q) 

U 

0 

k 

PI 
A digital cartridge tape recorder was also borrowed (from 

the underweter acoustics project) and used for on-line recording 

of the SCMP demodulates. This served as a back-up to the fieldtest analog tapes, and did not require a timing track and 

recorder calibration (a 7808 Hz timing track was recorded for the 

analog tapes, and tape calibration was required). 

Both the off-line and the real-time processing arrangements 

were first tested in our laboratory in Gaussian noise; then 

between different floors in the background noise of our 

laborztory, and finally in mine-to-surface transmissions in the 

field tests. 

4. Test Configuration - -- -------- 

- 4.1. Physical Site and Arrangement ---- - -- ---- 

The field tests occurred on Wednesday, June 29 and Thursday, 

June 33, 1383, at the Bureau of Mines Lake Lynn test-mine. This 

limestone mine which is near Uniontown, Pennsylvania, has an 

overburden of 100 meters at the test-position. The arrangement 

for the facility, the general test plan and configuration, and 

supervision of the data taking was the responsibility of the 

Bureau of Mines (handled by Project Officer Steven Shope). 

During the tests, S. Shope operated the transmitter and made all 

associated measurements. The analog tape recorder on the surface 

was operated by Bureau of Mines personnel (Czrl Gonoe) . The 

real-time processing equipment and digital recording apparatus 

was operated by Dr. K. Metzger of CEL. 

Figure 6 shows the general equipment arrangement used in the 

field tests. The underground loop antenna consisted of one turn 

of 300 feet (91.4 m) of 112 copper wire deployed in a rectangle 

resulting in a loop area of 2144 square feet (200 square meters). 

The current flowing in the loop was measured by the AC voltage 

drop across a small resistor placed in series with the loop on a 

Nagra tape recorder. The RMS current at the center frequency any 

time during the test could be obtained from the output of FFT 

computer, either directly at the mine or from the tape playback. 

- 4.2. Field-Test - Recording and Monitoring -4- -- -- 

The major data taking arrangements are also shown in Fig. 6. 

A Bureau of Mines (Collins) loop antenna fed a (U-M) pre-amp 

(Gain=666) , followed by a one-tenth octave (General Radio) filter 

centered at 1950 Hz. The input resistance of the pre-amp was 

about 208 kn , and a capacitor was plsced across the loop 

terminals to make the loaded antenna resonate at 1950 Hz. Also, 

some additional electrostatic shielding of the loop was 

accomplished by wrapping the loop with aluminum tspe. This 

reduced electrical noise pick-up in our noisy laboratory, and may 

have been helpful in the field. 
Digital 

Cartridge 

Recorder 

lOOm hes stone 

Overburden I 

Analog - O Tape 

NAGRA 

Analog 

Tape Recorder 

4V Wheat 

Bat 

r 

: - Pre- 1/10 

- me ' octave - 

Filter 

C r 

Fourier 

Tape Recorder Analyzer 

134' 

4V Wheat 

Bat 

1 

Figure 6. General Equipment Arrangement for Field Tests 

of Fig. 5 (c) 

, 

I 

I 

I 

. 

Transmitter 

LOOP 

i 

- - 

Rea 1-T ime 

Processing 
Two major dats-taking thrusts occurred during the field 

tests: 1) analog recording, for later processing, of the 

received signal on channel of an analog-reel tape, with a 7808 

Hz timing signal derived from a rubidium frequency standard 

recorded on a second channel, (this taping was done by the Bureau 

of Mines); 2) real-time processing and display for oneintegration time only (16 sequence periods = 4.46 minutes), plus 

d igital-cartr idge recording of the unaveraged demodulates for 

later processing. While the original field-test plan envisaged 

only the analog recording, we found we could use the DM12@0 to 

process the averaged signal in real time in the field to 

determine the threshold signal level, and at the same time 

provide a digital back-up for the analog tape recordings. 

The analog recording at the receiver site was done using a 

NAGRA IV-ST tape recorder supplied and operated by the Bureau of 

Mines. As already noted in Sec. 3.2, real-time processing 

averzged over 16 sequences was done at the site using the (CEL) 

DM1200, which calculated the demodulates and recorded the 

demodulates in real-time onto the digital cartridge tape. it 

also added the 16 periods and perfcrmed the matched filter 

operations on each 16 period signal sverage. rn avoid 

overloading the displays, the highest peak in each display line 

was normalized to unity. 

4.3. Test Series -- -- 

The data was taken on two contiguous days, Wednesday, June 

29, and Thursday, June 30, 1-983. Wednesd'ay morning was spent 

driving to the receiver site in a wooded area over the 

transmitter location. The receiving and recording equipment was 

set up in the back of a pickup truck and operated on AC power 

supplied by a gasoline engine-driven alternator. 

We chose to run each test for a period corresponding to the 

length of the analog tape reels. This permitted only two periods 

of the longest (17 minute) multiple integration times. The five 

data runs that will be described are listed in Table 1. 

On Wednesday, the noise background was relatively stable, 

consisting mainly of harmonic noise plus Gaussian. For the 

Thursday tests, the noise from harmonics of 60 Hz was the same as 

the previous day, but the noise from short random pulses and 

large noise bursts was at least 12 db higher than the previous 

day. We assume that the increase in short random pulses was due 

to atmospherics from a wide front of thunderstorms passing 

through the area. The large noise bursts appeared random to us. 

However, in playing back the tapes, we found that they consist of 

main noise bursts occurring every 45 seconds, each followed about 

6 seconds later by a second noise burst, usually lower in 

amplitude but longer in duration. Figure 1B (later) illustrates 

both the increased small pulse activity and the very large pulse 

pairs which were present on Thursday. 
TABLE 1. DESCRIPTION OF FIELD TEST DATA 

Tcs t 

Name Time 

C 3 : 00pm Wed. 

Antenna 

I 

35 min. 0.25 

I 

I 

35 min. 0.68 

35 min. 1.28 

I 

I Estimated Irlaut SNR (dB) 

1 . . Includes i Impulse Nolse Excluded Im~ulse Noise -- Based on I 

22Elz Rationale for 

, Siqnal Level Comments 

Wanted level high Normal background 

enough to permit noise; drop-outs 

on-site equipment occurred on time 

check-out, and reference due to 

later detailed poor quality tape 

diagnostics (such 

as phase behavior) 

Wanted significant 10 I1 

(20 dB) reduction 

from Test A 

Wanted repeat of Observed in field 

Wednesday probe that. noise increased 

for threshold; no over Wednesday; 

in-field detectio large bursts occurred 

Increase 6 dB, I# I) 

looking for field 

detection. None 

found . 

Increase another I I 11 

6 dB, looking for 

field detection. 

Signal was 

detected. 
5. Specific Signal Processin9 Methods Used I - -- ---- -- 

Figure 7 shows how the analog and digital playbacks were 

handled, using the SCMP program for matched filter averaging and 

time display outputs. For analog, the input to the DM1200 

computer was the output of the 1/18 octave filter and is the same 

signal as was recorded on the NAGRA recorder. The 7800 Hz sample 

clock signal is also the same as that recorded on the NAGRA. The 

signal input was sampled and converted at the 7800 Hz clock rate. 

The output of the demodulates consists of 4 inphase (x) 

demodulates and 4 quadrature (y) demodulates per sequence bit . The x and y sample outputs are interleaved. The keyboard 

controls whether these demodulates are recorded on the digital 

cartridge recorder, and whether averaging over 4, 16 or 64 

sequence periods was done before further processing. All of 

these operations were done in real time by the DM1200 computer 

using some handwired logic boards as well ss the SCMP program. 

The rest of the present SCMP program runs a little slower. 

For averages over 4 periods or more, it can process the averaged 

data in real time. Without averaging, it accepts new samples for 

on-line processing only when it is finished with the previous 

set. 

Notice that the signal to the display is normalized so that 

the lzrgest peak, whether signal or noise, is a full scale 

deflection. The CRT displsys up to 8 output pulse traces before 

erasing and starting over. Each output trace is identified by a 

trace number and the x coordinate position of the maximum peak. 

The phzse of the received signal at some specified x coordinate 

can also be printed on the display. This can be used to 

determine any frequency offset between the transmitter and the 

receiver.' Pressing a switch on the printer records the current 

CRT screen display on paper. 

All of the analog recorded signals as well as all digital 

recorded signals were processed using this system. The results 

chosen for this report (Figs. 11-14) are all from playback of 

digital recorded demodulates (lower left line in Fig. 7), except 

the clipper-processed ones, since we were not sure of the 

reliability of the analog recorded 7888 Hz sample clock signal. 

For the digital signals, the demodulates were formed during the 

field tests, and were recorded on the-digital tape. The effect 

of averaging over 1, 4, 16 or 64 times was investigated by 

running the same digital cartridge repeatedly with different 

processing commands. 2 

'~his feature was added to the program after the field test. 

The current program can also print out an estimated output SNR 

for each trace; this was not available during the data reduction 

for this contract. 

2~he data processing with clipping used the analog tape to 

generate a digital tape as shown in Fig. 9. 

Since the SCMP program was intended only for checking the 

data during the field test, there is no provision for converting 

the output to printed tables, or for recording it for further 

statistical analysis. 

The second digital processing program was used to print out 

tables of numerical values. This program, shown in Fig. 8, is an 

updated version of the program used in our previous work. It 

differs mainly in that the matched filtering was always done 

before averaging over additional periods. Also, the dat3 was 

recorded on 9-track reel-to-reel digit31 tape between processing 

steps. The demodulating and matched filter operations were done 

by the DM1200 using first the MTDEMOD and then the MTXC programs. 

The processed signal was recorded on the 9-track tape between the 

two st2ps. The matched filter output was again recorded on 9- 

track tape. Further averaging over a complete sequence bit, 2nd 

averaging over 1, 4, 16 and 64 periods was done at The University 

of Michigan Computing Center on the AMDAHL computer using a 

PRGG M5 program. This processing is essentially the same as that 

used and described in our previous work, Ref. 2, except that this 

time the matched filter operation was done by a second pass 

through the DM1200 computer using the MTXC program instead of 

using the AMDAHL. We did not run the output pulse displays on 

the AMDAHL because more completely labeled displays were already 

available from the SCMP program. 

The digit21 demodulates recorded during the field tests are 

an alternate source of demodulates which replace the MTDEMOD 

program (see dotted line, Fig. 5c). The numerical results quoted 

here are all from the digital tapes. The 7800 Hz sample clock 

was less reliable on the analog tapes. 

This program is much more flexible than the SCMP program, 

particularly since averaging operations are performed using 

Fortran programs running on the AMDAHL. However, th2 present 

AMDAHL programs, from PROG M1 through PROG M5, were all designed 

for displaying only the most recent 4 output pulses from each of 

the averaging procedures: xl, x4, x16, x64. This is convenient 

for zn operator display, but makes it difficult to compare the 

effect of different averaging times for signals obtained in 

fluctuating background noise levels. For example, a stray noise 

source may have been present during most of the time thzt the 16 

period and 64 period averages were formed, but turned off for 

part of the x4 and all of the last 4 xl pulses. Some of the x4 

and all 4 of the last xl pulses may then appear to have a much 

better SNR than expected from the noise observed in the longer 

averages. This is desirable for the operator display. However, 

with only 4 pulses available for each increment of averaging, the 

numerical output for real mine data, even when all the numbers 

are averaged, is too variable to be very useful for statistical 

analysis. The graphical displays from the SCMP program are more 

useful for comparing different averages than the numbers from the 

PROG M5 program. 

The tapes used on the NAGRA tape recorder for Tests A and C 

were of poor qua1i.t~ and had many dropouts on the time reference 

track. We constructed a phase lock loop to track through these. 

In general it was successful, but we have no assurence that the 

time track is free of errors. 

5.1 Off -Line MTXC Processing Arrangement ------ - 

Figure 7 shows the sequence of steps used in the off-line 

signal processing of both the analog tapes, and digital-cartridge 

demodulate-tapes. Analog tapes are digitized and demodulated in 

real-time by the DM1200 using the MTDEMOD program. The output is 

stored on tape DEMODF1. The digital cartridge tapes were those 

produced by the SCMP program. The MTXC program performs the 

first step (sequence removal) in the matched filter processing 

for both types of recordings off-line and records the 4 chips for 

each signal bit on digital tape XCF1. At the U-N Computing 

Center, the 4 complex chips are combined for the single period 

output. The 4, 16 and 64 period averages are formed on a sliding 

window basis. The MTXC output is also plotted on CRT and 

printer, but this is before the 4 chips are combined. 

5.2. Special -- Processing - for -- Displays 

In order to avoid preparing elaborate display programs for 

the off-line processing (above), we again exploited the DM1260 in 

the f~shion shown in Fig. 8, for the purpose of achieving display 

output. The SCMP program generates processed signal displays 

like those generated at the Lake Lynn Lzboratory during the field 

tests. The algorithm used for the matched filter is somewhat 

different from that used in MTXC. Also, the averaging over many 

periods is not done on a continuous sliding window basis, and 

therefore, the effect of normalizing the peak amp1 itude is 

slightly different. (It appears to us that this resulted in a 

more stable background noise level with the SCMP program). The 

displays of the test results (See Sec. 7) in this report are from 

this SCMP processing. A disadvantage is that, at present, the 

SCMP program does not generate a 9-track digital tape suitzble 

for sliding window processing and numerical tabulation at the U-M 

Computing Center. We therefore, have no numerical data for 

comparing the two algorithms. Cirect comparisons of the CRT or 

printer displays are not possible because the 4 complex chips are 

not combined in the MTXC process until the PROG M5 at the 

computing center. 

- 5.3. Special Clipper Signal Processing --- -- 

On Wednesday, more than one noise pulse occurred in most 

sequences, but the pulse amplitudes were comparable to the 

background noise. In contrast, the noise pulses on Thursday were 

both more numerous and much larger in amplitude. In addition to 

the isolated short noise pulses, there were bursts of periodic 

noise pulses, where, for many power line cycles, a large noise 

spike occurs in every cycle. The position of the spike shifts 
slowly in phase from cycle to cycle during a burst. The largest 

bursts occurred regularly every 45 seconds during the Thursday 

test. Each large burst was followed by a smaller but longer 

burst of spikes 6 seconds later (Fig. 10). These bursts were 

present through all of the Thursday tests until the last few 

moments of the last test. Then, their source apparently was 

suddenly turned off. 

Using the analog recordings, we did improve the Thursday 

results by subjecting the output of the input filter to a 

clipper. The pulse compression properties of the PN signal can 

be very effective in suppressing such noise pulses in a system 

designed for such conditions. (1) The signal duration should be 

such that only one (or s few pulses) is likely to occur in any 

one signal period. (2) Pulses much larger than the background 

noise should be clipped near the background noise level in a 

clipper with a bandwidth as wide as the noise pulses ahead of my 

narrow band filters. (3) A number of signal periods should be 

combined for reliable signal detection since some signal pulses 

may be reduced in amplitude by a long noise pulse or multiple 

pulses occurring in the same signal period. 

Since all our PN test data was recorded with the 1/10 octave 

filter at the front end, the second condition was not met. 

Nevertheless, a very significant improvement was obtained by 

clipping the NAGRA output near the peaks of the background noise 

level using the arrangement of Fig. 9. Under the conditions of 

Tests Q, P, and E, the signal has a constant amplitude more than 

40 dB below the background noise amplitude shown above. 

Clipping, therefore, has no effect unless an atmospheric noise 

burst is present. Then, the signal is largely suppressed during 

those times that the clipping levels are exceeded by the 

atmospheric burst. The signal segments that survive, however, 

still add coherently to form a (distorted) signal peak. 

6. Input SNR Aspects - - -- 

Input signal-to-noise ratio (SNR. ) is a suitable independent 1 

parameter for specifying the performance of a "detection" system, 

since: 1) the integration-time, 2nd hence, signal energy, are 

variable due to the parallel processing; and 2) we wish to 

compare the performance to that of the present system. 

- 6.1. Noise Types 

Our original system design exploited the simp1 ifying 

assumption that, if the major signal energy is contained between 

the relevant harmonics of the 60 Hz power system, the relevant 

background can be assumed to be stationary Gaussian noise. Also, 

the prior-effort laboratory demonstration tested the system in 

white Gaussian noise. It is known classically that, in Gaussian 

noise, with unknown signal phase (due to unknown onset and 

Moderate Pulse Noise 

on First Day (Test C) 

Heavy Duty Pulse Noise 

on Second Day (Test K) 

The Pulses from (b) Above with Vertical 

Scale Reduced to Show Complete Pulse Pairs 

Figure 10. Noise Sanples on First and Second Test Days 
propagation delay), the optimum signal processor is a matched 

filter, as implemented by our signal processing receiver. 

Both in preparing for and conducting the field tests, the 

signal was received us ins the Bureau of Mines suppl ied loop 

antenna. Both occasions provided us with experience of the 

effects of a (possible) combination of Gaussian noise, harmonic 

noise, and impulse noise. 

The harmonic noise appears to stem from periodic transients 

in the power distribution line voltage. The importmt feature of 

this noise is that it is periodic. 

The impulse noise may be due to either power line switching 

trznsients or atmospheric noise, usually from distant or loczl 

thunderstorms, that reach the antenna. If thunderstorms are 

within a few hundred miles, the noise becomes essentially nonstationary: heavy voltage surges are sporadiczlly imposed on top 

of the normally varying background noise. Hence, when 

atmospheric noise is not present, the background noise is 

relatively well-behaved and coarsely measurable; with strong 

atmospheric noise, measurements are quite difficult. 

For the t2st series described her?, on Thursday we 

encountered an unusually large impu1,se noise from an unexplained 

source. The large impulses occurred every 45 seconds, and the 

major pulse was always followed by a smaller pulse 6 seconds 

later. 

- 6.2. Input SNR Estimation -- - ---- 

In usual communication operations, one uses the iF bandwidth 

to define and measure the input nois2 to a communications 

receiver. In the test arrangement here, no specific filter 

served as an IF-equivalent. Rather, the signal passed through s 

1/10 octave filter, and the matched filter signal processing 

itself effectively filters the signal having a "main lobe" 

bandwidth. This was deemed superior to actually using a 30 Hz 

bandwidth filter centered at 195B Hz in the front end. In 

effect, the signal processing receiver implements a filter 

corresponding to a bandwidth of approxim3tely the reciprocal of 

the integrating time. 

The result is that the input SNR can be measured only 

coarsely both because: 1) the actual noise is constantly 

varying; and 2) the noise bandwidth which gives the most 

appropriate measure here is relqtively arbitrary. For this 

reason, three separate methods were used to estimate the SNR. 

As noted, both the PN system and the present puls? system 

are designed to operate in one of the minimum noise regions 

between the harmonics of 60 Hz. The input SNR must be measured 

in the same minimum and with the same bandwidth as the system for 

a meaningful measurement. For both pulse and PN receivers, the 
input bandwidth is usually taken to be the width of the main lobe 

of the (x Sinx) signal power spectrum. These bandwidths are 20 Hz 

for the pulse system and 38 Hz for the PN system. As a 

compromise in the first estimation method, the signal and the 

noise were each measured in a filter with a nominal 22 Hz 

bandwidth tuned to a center frequency of 1950 Hz. A small 

correction was 2pplied to estimate the noise in 20 Hz and 30 Hzwide bands. 

The noise power in this bandwidth was measured with true EMS 

reading voltmeters. Since we were interested both in the types 

of noise present and an estimate of the residual background noise 

after peak clipping, the noise power from the RMS voltmet~r 

output, taken from the analog t3pe, was recorded on a stripchart 

recorder or storage oscilloscope screen for each of the tests 

Fig 1). These results, for each of the data tests, appear on 

the left-most column of the SNR estimates of Table 1. 

A second estimate was based on a procedure suggested by the 

Bureau of Mines (S. Shope) . During prior and independent 

measurements at the same field-test site, and using the same 

transmitting antenna, the Eureau of Mines (Ref. 5) ~eesured the 

constant "Q" in the equation: 

INA HZ = - 

27rd 3 

where : 

I = antenna RMS current 

N = number of turns of antenna 

A = area of antenna loop 

d = length of transmission path 

Q = factor representing the signal loss due to the 

conducting half-space 

At 1950 Hz, the Q was determined to be .dg9. 

If we had some way to estimate the H coming from the noisc- z 

only measurements, we could obtain a theoretically-predicted 

estimate of the surface SNR during any giv~n integration period. 

The Eureau of Mines has a method for computing an HZ from noise 

measurements (see Appendix E). For our purposes, we used a 

theoretical HZ value (-074 A/m) based on a minimum noise 

experienced at EI previous mine which was not operating (Ref. 6). 

In any event, it may be true that surface noise at a non- 
operating mine may be coarsely consistent at different locations 

and at different times. The numbers shown in the middle SNR 

column of Table 1 resulted from this exercise. Note that both 

this theoretical predication and the previous 22 Hz measured SNR 

do not include the large impulse noise. 

The third method of estimating input SNR consisted of 

examining the output peak behavior relative to the intervening 

noise-only behavior, which now includes the impulse noise. The 

output geak-pulse value is interpreted as E +n b 0' and the 

intervening values are interpreted as ng. Then, using the 

classical relation: 

Eb/nO = (S/K)i x TW = (S/N) x (sequence length) 

and 

A hand-calculated method was used to perform this estimate. 

The right-most column of SNR numbers in Table 1 resulted from 

this calculation. Examples of the output for this estimate 

appear in Appendix A. Appendix C contains the actual signal 

levels and noise levels that were used in the SNR estimates. 

The following comments are intended to document what we 

learned about the noise behavior during these tests. Curing 

calibrating tests M, (PN), and L, (pulse), which were taken prior 

to the data tests on Thursday, the transmitting antenna currents 

were high enough so that the received signal volt?ges in the 22 

Hz filter band measurement could be measured directly. The 

unmodified Collins antenna output was connected directly to the 

NAGRA recorder input for these tests. On playback, the NAGRA 

output was fed to the 22 Hz bandwidth filter and the output 

measured with the true RMS voltmeters. The signal voltage at the 

Collins antenna was then calculated from the known NAGRA gain 

settings. 

In calibration test J, taken during the same period, the 

same system was used (with higher NAGRA gain settings) to record 

the background noise level. The noise power read by the true RMS 

voltmeters was recorded on a stripchart recorder and on a storage 

oscilloscope screen. The background noise level was estimated 

from these displays. It was assmed, as in the left part of the 

table, that the large and very unpredictable noise pulses would 

be removed by peak clippers in an operational system. 

Considering the crude nature of the theoretical predict ions, 

the directly measured input SNR and the backward-calculated one 
appear consistent with the theoretical predictions. Note that 

from the measured Collins antenna voltage at this high 

trznsmitter current, one could calculate the Collins voltage for 

each of the other Test transmitter currents. This could be used 

to czlculate a set of input SNR for the Collins antenna and the 

background noise of Thursday. 

This is essentially what was done for the PN tests listed at 

the right edge of the table. However, the background noise 

voltage was the noise voltage measured at the output of the 

modified resonant Collins antenna used with the PN receiver, and 

the signal voltage was obtained from the strongest signals 

receivea with the modified mtenna (Tests K, 0, A, and C) . Again, the input SNR for each test appears compatible with that 

in the other sections of this chart. Again, we would expect all 

of the tests signals (with clipping) to be detectable. 

We note that the SNR does not depend on either the mtenna 

gains or the systzm calibration as long as the same system is 

used for both the signal measurements and the noise measur2ments. 

There is general agreement in these values. However, the actual 

voltages for our resonant antenna system are roughly 30 times 

those expected for the theoretical non-resonant antennz used in 

the predictions at the right of the table. This appears 

consistent with some simple measurements we have made on the 

resonant antenna, but we have no absolute calibrstion for the 

antenna which we could use, for example, to determine the ground 

conductivity from the measured antenna voltages. The voltages we 

obtzinea for the unmodified Collins antenna are also higher than 

those predicted for a non-resonant system. This also appears 

consistent with our experience. Before we tunea our antenna, it 

was self-resonant roughly 3 KHz above our operating frequency. 

These points would have to be checked if the PN system were to be 

used for ground conductivity measurements. Fortunately, they 

have no direct effect on the communication system SNR 

calculations. 

7. Test Results - - ----- 

After processing the tapes, it was found that the maximum 

sensitivity for the processing-receiver occurred for 16-sequencelong integration (4.46 minutes), and not for the expected 64- 

sequence length (17.854 minutes) . Using the high signal levels 

in Test A, we were able to show that this result was 5ue to some 

drift between the transmitting oscillator and the receiving 

oscillator. We do not know the reason for this drift, but the 

result is that, for these field tests, the optimum results occur 

when one limits the integrating time to 4.46 minutes. This is 

probably the maximum tolerable delay time in an operational 

system; nevertheless, for purposes of demonstr3ting maximum 

signal processing capabil ity, we did or iginally intend to 

coherently integrate up to 17.854 minutes. 
For Test A, (Fig. lla), and Test C (Fig. llb), the signals 

are detectable with 1, 4, 15 or 64 sequ2nce additions. Hence, 

the earliest alarm in this case would haw occurred in 16.54 

seconds. Figure lla shows six samples of the receiver output 

time axis when the minimum integr'ation time (16.74 seconds) is 

used. The input SNR is estimated to be 28.4 dB. As seen in 

Fig. lla, the output time axis shows no off-pulse activity, which 

means that the output noise is lower than the smallest digital 

quantization interval. The significance of this is that the 

input SNR is much larger than required. The data reduction 

program computed the received phase angle (A) for the time 

position identified as the maximum (reference) posit ion. In 

Fig. lla, we see that the phase angle changed by 15' in 1.4 

minutes. This indicates a transmitter frequency error of 2.6 

7 parts in 18 . Also, 2s seen, the output noise is at the 

resolution limit of the display (or lower) . 

Figure llb shows the results of Test C, where the input SNR 

is estimated to be 2.14 dB, with integration time still equal to 

the (minimum) 16.74 seconds. Hence, again the alarm would have 

occurred 16.74 seconds after transmitter turn-on. Again, the 

input SNR is sufficiently high so that off peak output "noiseonly" values are below the quantization level of the processor. 

Figure 12a shows a series of six output time axes, from Test 

Q, when the input SNR was estimated to be about -11.7 dB, which 

included the erratic thunderstorm impulsgs and the large periodic 

impulses. The integration time is now 4.46 minutes, the longest 

effective integration time for these field tests (recall, the 

17.854 minute integraton was less effective because of frequency 

drift between oscillators) . We us4 the of £-line clipping 

arrangement of Fig. 9 to achieve the clean results of Fig. 12a. 

Figure 12b shows the same data, with same integration, but 

without use of clipping. 

Figure 13 shows a series of 10 output time axes for Test P, 

where the input SNR (with impulse noise excluded) was estimated 

to be about -19.9 dB. The integration time was again 4.46 

minutes and clipping was required. We estimate that, for alarm 

purposes, the SNR is still somewhat above threshold for reliable 

detection with negligible false alzrms. 

Figure 14 shows a series of 10 output time axes for Test E, 

where the input SNR (with impulse noise excluded) was estimated 

to be -38.8 dB. The integration time was 4.46 minutes, and 

clipping was required. We believe this test coarsely corresponds 

to threshold operation for a reliable alarm system. 

- 8. Conclusion and Recommendations -- - 

The major conclusion is that the field test described herein 

definitely demonstrated in the field the viability and potential 

0' 

-4 

(I] 

V: 

0, 

U 

0 

a 

C 

-4 

a 

(I] 

3 

0' 

.PI 

a 

a 

-4 

rl 

U 

0 

z 

h 

P 

Y 


of the PN-signal plus coherent-inteqrqtion technique that we are 

pursuing for the Bureau of Mines. We demonstrated, using 

integration times of 4.46 minutes, reliable operation at 

transmitter antenna currents of 8.25 mA, corresponding to about 

an input SNR of -38.8 dB. 

The superiority of the PN signal was demonstrated during 

these field tests. One elementary but crucial aspect is the 

(relative) inherent detectabil ity of the output "pulse" that 

results from matched-£ ilter ing a signal having an ideal (pulse) 

autocorrelation function. Another superiority of the PN 

signal-plus-stable clock results from the true-zero that occurs 

in the power spectral density at harmonics of the 60 Hz power 

system. The true zero apparently is reliable and stable because 

the large power-distribution system has a stable and wellcon trolled frequency. The stable oscillator used in our system, 

for coherent integration, in combimt ion with the stable powerfrequency, permits a suppression of harmonics which is dif f icu? t 

to match with steep-sloped filters. 

Still another advantage of the PN signal here is its 

compatibility with extending the alarm-function discussed thus 

far to ensuing uplink communications. The pulse-output is 

sensibly compatible with a high M-ary pulse-position modul.ation. 

The achievable uplink data rate is proportional to loq2M and 

inversely proportional to the total integration time required. 

This is discussed further later when considering recommendat ions. 

The attractiveness of parallel processing, which implements 

an adapt ive-type processing, was demonstrated during these tests. 

In effect, the alarm occurs at the earliest time, commensurate 

with the input SNR. Operational design (say) would probably aim 

to have the alarm occur at the minimum time (say 15 seconds). 

Then, the longer signal-processing times simply guard those 

conditions under which some unexpected phenomenon results in a 

lower SNR. Although elementary in concept, t.his appears to be 

thoroughly sensible for providing a reliable alarm and 

communication system. 

As usually occurs, the field tests here revealed certain 

improvements thet are sensible and warrmted. First, we now 

realize that operation will be improved by using a clipper ahead 

of any filtering , which is adjusted to somewhat -3bove the 

"between 60 Hz pulses" level of the ongoing time axis. Such 

clipping will also do the best job possible of reducing the 

effect of atmospheric pulse noise. We would have experienced 

somewhat improved results from this field test if such a clipper 

had been used. 

Another improvement would have been available if the 

frequencies had not drifted somewhat over the maximum planned 

interval of 17.84 minutes. Although we could establish, by 

detailed diagnostic examination of the data during the high SNR 
tests, that the drift occurred, we do not know the cause. This 

aspect will warrant further attention in moving toward an 

operational system. 

It is the authors' opinion that one significant application 

of the new system lies in using fixed-point, round-the-clock 

receiving sites to replace the initial detection now done by 

heliocopter-search followed by walking over the indicated ar3a. 

Another significant application lies in using a localized nullseeking strategy to posit ion-locate the transmitter. 

Having now demonstrated the feasibility of an ultra-rzii3ble 

signalling system, it is recommended that the system be refined 

and an implementation strategy be developed. Suggested 

refinements include: 

a) Design and develop a surface "base station," that 

efficiently accomplishes the signal processing. 

b) Investigate the possibility of uplink voice, using some 

low-bit-rste voice digitization, plus an efficient 

digital modulation. 

c) Investigate posit ion-location improvements znd 

techniques that are offered with this signalling 

approach. 

The implementation strategy should include consideration of 

general communication or measurement opportunitiea that are 

opened up by this system. 
APPENDIX A 

A Sample of the Printed Data Cutput of the EN Processor 

Ervin Holland-Moritz 

All of the analog recorded PN test data as well as the 

digital recorded PN data was processed and printed out in tabular 

form for detailed analysis. Including clipping tests, there are 

more than 25 sets of such printed tables, each consisting of more 

than 130 pages with 8 columns of data per page. Part of a sample 

page from the Test A output is shown in Table A-1. This sample 

is from the digital recording and was processed with no clipping 

or low frequency noise rejection. 

The index column (1) on the left represents increasing timz 

from start of processing in intervals of 8.2 milliseconds. In 

each time interval a new processed output is computed from the 

demodulates and printed in the following column. If any signal 

pulses are present, they will be separated by 2040 in the index 

number count, but the signal may start at any index number. 

Columns (2) and (3) are processed outputs with coherent 

integration over only one FN sequence period. Column (2) 

consists alternately of the processed inphase demodulates and the 

processed quadrature demodulates. There are 4 pairs of 

demodulates per PN signal chip. Prior to index #901 and after 

#9.1E, numbers represent alternately the inphase and quadrature 

noise voltages. In between, the voltages are a combination of 

noise and signal. The signal peak occurs at index #903. If the 

SNR is adequate, the received signal phase can be determined from 

the magnitudes of the adjacent inphase and quadrature voltages. 

Note that there are still 4 processed demodulates per signal 

pulse at this point. In the final processing, a coherent sliding 

window average over 4 pairs of demodulates is taken. 

The processed output from column (2) is printed in column 

(3). The inphase processed demodulates are combined in a sliding 

window sum over 4 and the window output is squared. The 

quadrzture demodulates are similarly combined and squared. The 

number printed in column (3) is the sum of the squares of the 

inphase and quadrature components. Note that the index number 

and sliding windows then shift by one demodulate so that each 

processed demodulate is used twice, once with its preceding 

quadrature component and once with its following quadrature 

component. Since these output numbers are based on samples taken 

at fixed time intervals, they can be interpreted either as 

(signal+noise) power or (signal+noise) energy. In particular, 

the peak number is the energy of the PN signal (with noise) after 

coherent integration over one sequence period. 

Similarly, column (4) consists of processed demodulates 
O"@? 

:r 

NCJ:%Q, ut-.CIm 

6E)TN 

r-nv,m 

-J(Ca.-F: 

- c 6.7 L'I ,r: L-l aD --- 

u' u3 0 0 

r-i rl 

X X u'rl 0 ... 

dd 0 0In d rlm m II II I1 m >z 

R \ 

z m 
after coherent integration over 4 PN sequence periods. Each 

demodulate in this column is obtained from the average of 4 

correspon6ing demodulates from 4 consecutive pulses in the column 

(2) data. Column (5) is the processed output after coherent 

integration over 4 PN sequence periods. It is obtained from 

column (4) by the same sliding window and squaring process 

described above for column (3). 

Column (7) is the output after integration over 16 periods, 

while column (9) is the output after integrstion of 64 periods. 

In each case, the new processed demodulates are obtained by 

averaging the corresponding processed demodulates from 4 PN 

sequence periods from shorter integration time. 

Because the new demodulates are obtained as an average, the 

ideal pulse outputs for a noise-free system with ides1 

integration of a constant signal should remain the same with 

increasing integration time. If the noise is uncorrslated with 

the signal (for example, in Gaussian noise), then the average 

noise power should decrease by 6 dB in going from one integration 

time to the next longer time. The detailed printed data can be 

used to determine if the system, operating in real mine noise, is 

performing as designed, and for finding improved processing 

algorithms for increasing the reliability of signal detection. 

However, this output is much too detailed for display to an 

operator in real time. Graphical pulse displays, obtained either 

by plotting these numerical tables (or obtained more directly 

from the SCMP processing sequence), are more useful in evaluating 

performance under operational conditions. 

Note that in the present numerical processinq, only the oate 

for the most recent 4 pulses for each level of integration is 

saved in the computer arrays. The earlier data was used in 

calculating the longer term averages, but at a later time cannot 

be displayed separately without rereading the MTXC data tape. 

Since there are at least-128 PN pulse sequences in a single test, 

printing a complete list of all the data pulses in this detailed 

manner is impractical, especially in an operations1 system. On 

the other hand, it is quite practical to display 128 consecutive 

pulses on a graphical printer output. 

The numbers printed at the bottom of Table A-1 are the 

computed signal powers and the average of the noise power samples 

for the particular pulses shown. Eecause the times of 

integration increase towerds the right, the longer integration 

times may include large noise pulses which may, or may not have 

been present in the shorter data periods displayed towards the 

left. 

We see that the coherent signal integration is quite good up 

to integration over 16 periods. Then the drift in transmitter 

frequency caused a significant loss in signal power. The average 

noise power varies widely from pulse to pulse because of the 

noise statistics. In some cases, it decreases at more than the 
expected 6 dE in going from one integration time to the next. In 

other cases, the decrease is less than 6 dB. Any consistent 

decrezse of less than 6 dB indicates that there is some 

correlation between the existing noise and the signal (for 

example, 60 Hz harmonics in the signal band). In principle, such 

correlated noise can be removed by improved processing 

algorithms. 
APPENDIX B 

Comparison of the Pulsed System to the PN Method 

Steven Shope 

U. S. Bureau of Mines 

An objective of the June field test was to evaluate the 

performance of the PN communication method as a trapped miner 

location technique. The performance of the existing pulsed 

location system has been well defined through earlier research 

efforts. A rough approximation of the PN system's performance 

can be made by field testing the two systems at the same site 

under similar noise conditions. It must be realized that this 

compsrison serves only as a crude estimate and is by no means 

conclusive. 

At the Lake Lynn test site, the 1958 Hz pulsed trapped miner 

transmitter was chosen as the reference device. The underground 

antenna configuration was the same for both transmitters (pulsed 

and PN). The signals on the surface were tape recorded for later 

analysis. The current in the underground antenna was calculated 

by measuring the voltage drop across a shunt resistance placed in 

series with the antenna loop. An averaging FFT was used to make 

this voltage measurement. 

Analysis of the surface vertical magnetic field revealed the 

pulsed transmitter produced 37.27 dB (re: 1 amp/m) . The vertical 

magnetic noise field excluding 66 Hz harmonics was 4.60 dB 

(re:l amp/m). The net SNR on the surface due to the pulsed 

transmitter was 32.67 dB. The underground magnetic moment was 

2 105 ~m~-m~ (-526 amps flowing in a loop of area 200 m ). For the 

sake of argument, say the minimum detectable SNP (for some 

probability of detection and false alarm rate) is 10.2 dB. Since 

the system is linear, the pulsed transmitter must produce at 

least .396~18-~ amps (-526 amps down by (32.67-18.2)dB) for the 

signal to be detected on the surface. In other words, .?96~18-~ 

amps would produce the 10.2 dB minimum SNR on the surface. 

In test "E" of the PN technique, 250 amps were flowing in 

the loop. On the surface, after detection and processing, the 

receiver produced a 18.2 SNR output. Thus, we have similar 

signals from both systems, yet the PN method required only 258 

amps while the pulse system required .396~10-~ amps. The 

difference in the two systems is 44.0 dB. This figure may be 

roughly used as the performance improvement one obtains by using 

the PN technique. 
APPENDIX C 

Input SNR Estimates 

Ervin Holland-Moritz 

Table C-1 contains the detailed numbers from which the input 

noise estimates of Table 1 (in text) were obtained. In addition, 

the table contains an additional set of "expectedI1 noise and 

signal values. The "expected" field strength and the resultant 

antenna voltages were calculated from the equations in Table C-1, 

using Q=.409 from previous Bureau of Mines measurements at this 

site. The voltage equation assumes 3 non-resonant mtenna coil. 

Measurements on the antenna coils indicate that both coils 

actually are resonant and this increases the antenna output 

voltage. The Collins antenna is self-resonant at a frequency 

considerably higher than the signal frequency. The exact 

resonant frequency is determined by the input impedence of the 

measuring device and the cable capacitance. The U of M antenna 

was tuned to be resonant at the signal frequency. 

The expected noise used was taken as the "minimum" noise 

experienced in pr ior noise tests (Westinghouse Final Report) , which were used by Collins in their receiver antenna design. 

These noise voltages are measured by the resonant antenna coils 

in the same proportion as the signal voltages. The calculated 

signal to noise ratios, therefore, are not affected by antenna 

resonances. 

The total received noise was much higher on the second day, 

but the increase was largely due to impulse noise. The noise 

levels shown in Table C-1 were obtained from measurements of 

chaxt recording or storage oscilloscope images where the noise 

pulses were ignored and the continuous background noise was 

measured. The noise measurements for the second day, therefore, 

correspond to the expected noise levels after clipping . 

Calculated in this way, the calculated SNR's for the 

processed PN signal appear consistent with the visual appearance 

of the corresponding processed time axis displays. 
TABLE C-1. RECEIVER INPUT SIGNAL AND NOISE CALCULATIONS 

I 1 :HZi 'EX~ 

An tcnna Signal 

Day ,'I'ca>;!_ Current lrA/M I Voltaqe ----- 

:ted Antc 

5~40 isc 

7- 

Vol taqe 

Mine I 

inimum" 

~na Measured Collins Antenna Measured U of M Antenna 

Input 4~ignal 6~oise Input ' Signal 6~oise Input 

S/N -Val tage VoLtage S/N Voltage IVoltage S/N - 

~ise - AC 

I 

Power for 

I 

Receivers Turned 

L ' 

Off 

-1-1,- 

Measured Collins Antenna I Measured U of M Antenna 

NAQ 2 1. Ref. A Hz - [2rdl] x I A = 200 M , Q = 0.409, d = 100 M, N = 1 

2. Ref.6 V = L8n2 ;; 1~~ x Ax F x N] x Hz = .87*H, A = 0.113 M2, f = 1950 Hz, N= 500 

3. khlS during 1/10 second pulse 

4. RMS averaged over 1 second/pulse 

5. Calculated from minimum mine noise Hz = .a74 $ , Ref. 6 

S?!R Y(51~;urd-I f ram 

Output PrintOuts (Includes 

Impulse Noise) --. ----- --. -- ---- 

6. Steady noise component only, measured in 22 Hz filter bandwidth. Does not include impulse noise. 
REFERENCES 

1. M. P. Ristenbatt, "Post-Disaster Communication CesignConcept," Final Report for Contract No. G0177073, Cooley 

Electronics Laboratory, The University of Michigan, Ann 

Arbor, Michigan, February 1979. 

2. M. P. Ristenbatt and E. K. Holland-Moritz, "Evaluation 2nd 

Demonstrat-ion of a New Post-Disaster EM Communication 

Technique for Mines," Final Report for Contract No. 38199109, 

Cooley Electronics Laboratory, The University of Michigsn, 

Ann Arbor, Michigan, 19P1. 

3. Simmons, "Development and Prototype Production of a Trapped 

Miner Signalling Transmitter/Trznsceivzr," Contract 

No. J0395@17, General Instrument, Government Systems 

Division, Hicksville, New Yor k, June 19E1. 

4. Kurt Metzger, Jr., "Signal Processing Equipment and 

Techniques for Use in Measuring Ocean Acoustic Multipath 

Structures," Doctoral Cissertation, The University of 

Michigan, Ann Arbor, Michigan, 1983. 

5. Steven Shope, "Lake Lynn Field Test Site Electrical 

Characterization," Memo to M. P. Ristenbatt, July 1953. 

6. Rockwell International, Collins Commercial Telecommunications 

Division, "Waveform Generator-Package and Receiver," 

U.S. Department of the Interior, Bureau of Mines, Contract 

No. Ha242810 Final Report, November 1976, pp. 20 and 41.