TECHNICAL SERVICES FOR MINE 

COMMUNICATIONS RESEARCH 

TRANSMIT ANTENNAS FOR PORTABLE 

VLF TO MF WIRELESS MINE COMMUNICATIONS 

Robert L. Lagace - Task Leader 

David A. Curtis, John D. Foulkes, John L. Rothery 

UNITED STATES 

DEPARTMENT OF THE INTERIOR 

BUREAU OF MINES 

USBM CONTRACT FINAL REPORT (H0346045) 

Task C, Task Order No. 1 

May 1977 

ARTHUR D. LITTLE, INC. 

Cambridge, Massachusetts 

Arthur D Little, Inc. 
REPORT DOCUMENTATION PAGE 

1. Report No. I 3. Recipient's Accession No. 

4. Title and Subtitle 5. Report Date 

Technical Services for Mine Communications Research May 1977 

TASK C(T.O.1) 6. 

TRANSMIT ANTENNAS FOR PORTABLE VLF TO ME WIRELESS MINE 

Tf'ATTnNS 

7. Author(s) 8. Performing Organization Report No. 

Robert L. Lagace, David A. Curtis, John D. Foulkes, 

John L. Rothery ADL-77229-Task C 

9. Performing Organization Name and Address 

Arthur D. Little, Inc. 

Acorn Park 

Cambridge, Massachusetts 02140 

I 10. Project/Task/Work Unit No. 

11. Contract or Grant No. 

H0346045-Task Order No. 1 

13. Type of Report 

12. Sponsoring Organization Name and Address Final (Task C) Office of the Assistant Director-Mining May 1974 - May 1977 

Bureau of Mines 

Department of the Interior 

Washington, D. C. 20241 14. 

15. Supplementary Notes 

1 16. Abstract 

An investigation is made of the feasibility of developing compact transmit antennas and/or 

other means to efficiently couple VLF to MF band radio energy between portable wireless 

communication units in coal mines. The completely wireless communication ranges between 

two portable radios equipped with practically-sized reference loop antennas in representative coal mine environments are estimated. Antenna technology is assessed with respect 

to transmit moment, range, intrinsic safety, battery and wearability requirements to determine the most suitable antenna types for use by miners. It is concluded that air-core 

bandolier loop antennas, and erhaps ferrite-loaded loop antennas, having peak transmit 

moments of about 2.5 ampere-m5 will be the most effective antenna choices for portable 

units. This choice, however, will not allow the range goal of 1350 feet to be attained in 

mines having unfavorable signal attenuation and noise environments. Rectangular air-core 

loops are compared with grounded horizontal-wire and vertical-wire mode exciters for fixed 

station applications. The potential for significantly increasing the wireless communication range by inductively coupling to communications and power lines in a mine is evaluated and found to be considerable based on theoretical coupling equations. Lastly, the 

theoretical performance of a hybrid noise cancelling diversity receiving technique that 

utilizes both the electric and magnetic field components of the ambient noise to improve 

the receiver output signal-to-noise ratio under certain adverse conditions is estimated 

and shown to be favorable when the magnetic and electric field noise components are highly 

correlated. 

17. Originator's Key Words 

Mines - Radio Communication 

Antennas - VLF to MF 

Radio Propagat ion - Conducting Media 

Noise-Diversity Cancelling 

Coal Mine Communications 

18. Availability Statement 

19. U.S. Security Classif. of the Report 20. U.S. Security Classif. of This Page 21. No. of Pages 

157 

22. Price 
USBM H0346045 

TECHNICAL SERVICES FOR MINE 

COMMUNICATIONS RESEARCH 

TRANSMIT ANTENNAS 

FOR PORTABLE VLF TO ME' WIRELESS MINE COMMUNICATIONS 

Robert L. Lagace -- Task Leader 

David A. Curtis, John D. Foulkes, John L. Rothery 

ARTHUR D. LITTLE, INC. 

Cambridge, Massachusetts 02140 

C-77229 

The views and conclusions contained in this document are 

those of the authors and should not be interpreted as 

necessarily representing the official policies or 

recommendations of the Interior ~epartment's Bureau of 

Mines or of the U. S. Government. 

USBM CONTRACT FINAL REPORT (H0346045) 

Task C, Task Order No. 1 

MAY 1977 

UNITED STATES 

DEPARTMENT OF THE INTERIOR 

BUREAU OF MINES 

Arthur I> Little, Inc 
FOREWORD 

This report was prepared by Arthur D. Little, Inc., Cambridge, Massachusetts under USBM Contract No. H0346045. The contract was initiated 

under the Coal Mine Health and Safety Program. It was administered under 

the technical direction of the Pittsburgh Mining and Safety Research 

Center with Mr. Howard E. Parkinson acting as the technical project 

officer. Mr. Michael W. College was the contract administrator for the 

Bureau of Mines. 

This report is a summary of the work recently completed as part of this 

contract during the period May 1974 to May 1977. This report was 

submitted by the authors in May 1977. 

No inventions or patents were developed and no applications for inventions or patents are pending. 

iii 

Arthur I> I-ittle Inc. 
TABLE OF CONTENTS 

Page 

List of Tables 

List of Figures 

I. EXECUTIVE SUMMARY 

A. OBJECTIVE 

B. APPROACH 

C. FINDINGS AND CONCLUSIONS 

1. Wireless Range Estimates 

2. Antenna Technology 

3. Coupling to Mine Conductors 

4. Noise Cancelling Diversity Reception 

11 MINE WIRELESS RADIO RANGE ESTIMATES 

A. SUMMARY 

B . INTRODUCTION 

C. THE HOMOGENEOUS MEDIUM MODEL 

1. Signal Field Strengths 

2. Range Estimates 

a. Single Sideband Voice 

b . Narrowband Paging 

D. THE THREE-LAYER MODEL 

1. Signal Field Strengths 

2. Range Estimates for FM Voice 

3. Discussion of Implications 

111. ANTENNA TECHNOLOGY 

A. SUMMARY 

viii 

Arthur D Little Inc. 
TABLE OF CONTENTS (Continued) 

Page 

B. ELECTRICALLY SMALL ANTENNAS 

Electrical Smallness 

Radiation Resistance 

Efficiency 

Quality Factor-Q 

Power Factor 

Effective Volume 

Application Implications 

Impact of Conducting Media 

C. PORTABLE MANPACK ANTENNAS 

1. Conventional Whip Antennas 

2. Active Whip Antennas 

3. Conventional Loop Antennas 

4. Ferrite-Loaded Loops 

5. Special Multi-Turn Loop (MTL) Antennas 

D. FIXED STATION ANTENNAS 

1. Vertical Plane Loops 

2. Alternative Mode Exciters 

a. Horizontal Grounded Long Wires 

b. Vertical Grounded Wires 

3. Other Loop Antennas 

E. REFERENCES 

IV. COUPLING TO EXISTING CONDUCTORS IN MINES 

A. SUMMARY 

B . INTRODUCTION 

C. REPRESENTATIVE GEOMETRIES 
TABLE OF CONTENTS (Continued) 

Page 

D. COUPLING TO SIMPLE TWO-WIRE LINES 

1. Case la - Coplanar Geometry 

a. Electric Field Method 

b. Magnetic Field Method 

2. Case lb - Aligned Geometry, Magnetic Field 

Method 

3. Unfavorable Geometries 

E. COUPLING TO TWISTED PAIR CABLE 

F. LOOP-TO-LOOP COUPLING VIA THE LINE 

G. SIGNAL-TO-NOISE RATIO ESTIMATES 

V. HYBRID NOISE CANCELLING DIVERSITY RECEIVING TECHNIQUE 

A.. SUMMARY 

B . INTRODUCTION 

C. SIGNAL-TO-NOISE RATIO IMPROVEMENT ANALYSIS 

1. Nomenclature 

2. Signal-to-Noise Ratio 

3. Circle Diagram Interpretations 

4. Expected Values 

5. Boundary Plots and Performance 

D. RECOMMENDED FIELD MEASUREMENTS 

vii 

Arthur D Little, Inc. 
LIST OF TABLES 

Table No. 

2-1 Range Estimates for Wireless Communication 

with Personal Portable FM Radios in a HighCoal Seam 

Page 

Typical Dimensions for Structures of Interest 

in a Mine 96 

Regions of Validity for Using BR <<1 Criterion, 

the Induct ive-Field Approximat ion (In Air) 101 

Values of Eg in the Plane of a Loop at a Distance 

R Away in Air 10 7 

Currents in uamperes Induced in Conductors for 

the Situation of Figure 4-la (In Air) 108 

Voltages Induced in Cable Conductors in the 

Situation of Figure 4-lb (In Air) 

viii 

1- 

Arthur D Little Inc. 
LIST OF FIGURES 

Figure No. Page 

2-1 Magnetic Field Strength /H / from a Loop Antenna 

Z in an Air Medium 10 

Magnetic Field Strength /H / from a Loop Antenna 

in a Conducting Medium andZRMs Magnetic Field 

Noise in Mines (Normalized to 2.5kHz Bandwidth) 11 

Comparison of Loop Signal in Conducting Medium 

(from Figure 2-2) and Mine Noise Near Operating 

Machinery (Normalized to 2.5kHz Bandwidth) 13 

Comparison of Loop Signal in Conducting Medium 

(from Figure 2-2) and Mine Noise in Haulageway 

(Normalized to 2.5kHz Bandwidth) 

Magnetic Field Strength /H / from a Loop Antenna 

in a Conducting Medium andZRMs Magnetic Field 

Noise in Mines (Normalized to a lOHz Bandwidth) 

Three Layer Model Geometry for Low and Medium 

Frequency Radio Wave Propagation 

Theoretical Magnetic Field Strength Plots Versus 

Distance and Frequency for Three Layer Model 

for Two Values of Coal Conductivity 

o =1.4 x 10-4 Mho/m and o Mho/m 

C 

20 

C 

Representative RMS Magnetic Field Noise Levels 

Measured in the McElroy Coal Mine 23 

Representative RMS Magnetic Field Noise Levels 

Measured in the Robena and Itmann Coal Mines 24 

Behavior of Antenna Intrinsic Q (Based on Chu (1)) 35 

Effective Volumes and Radiation Power Factors 

for Electrically Small Antennas 

Illustration of Radiation Power Factor Limitations 

Imposed by Antenna Effective Volume Relative to 

Volume of a Radian Sphere 39 

Comparative Examples of the Effective Volumes 

of Several Common Antenna Types 40 

Arthur D Little, lnc 
LIST OF FIGURES (Continued) 

Figure No. 

3- 5 

3- 6 

Page 

Example of an LF Band T-Antenna 43 

Example of State-of-the Art Large VLF Antenna 

in Northwest Cape Australia 

Active Transmit Antenna FEBL Configuration 

FEBL Configuration for Maximizing Antenna 

Current Moment 

Equivalent Circuits 

Sketches of Several Bandolier Loop Antenna 

Configurations 

Minimum Igniting Current as a Function of an 

Inductive Circuit for an 8.3% ~ethane/Air 

Mixture 

Minimum Ignition Voltage as a Function of 

Capacitance for Ignition of an 8.3% Methane/ 

Air Mixture 

Typical Configuration for Loops Loaded with 

Ferromagnetic Cores 

Sketch of Flux Concentration Caused by High 

Permeability Rod 

Apparent Permeability, prod, as a Function of 

Rod R/d and Rod Intrinsic Permeability, v 

M'TL Antenna Diagram and Impedance Characteristics 

Wall-Mounted Fixed Station Loop in Coal Mine 

Tunnel 

Fixed-Station Turnstyle Loop Antenna Configurat ion 

Turnstyle Antenna Installation Possibilities in 

a Coal Mine Environment 

Propagation at ELF in the Earth-Ionosphere 

Wave guide 

m 

Arthur D Little Inc. 
LIST OF FIGURES (Continued) 

Figure No. 

3-21 Sketch of a Grounded Long Wire Antenna 

Installed in a Coal Mine Tunnel 

3-22 Long Wire Turnstyle Configuration in an ~ntry/ 

Crosscut Tntersection 

3-23 Long Wire Array Configurations in a Mine Entry 

and a Crosscut 

3-24 Sketch of a Grounded Vertical Wire Antenna 

Installed in a Coal Mine Tunnel 

3-25 Conceptual Sketches of Vertical Wire Mode 

Exciter Configurations 

4- 1 Different Geometries for a Horizontal Loop 

Antenna and a Twin-Lead Cable 

4-2 Different Geometries for a Horizontal Loop 

Antenna and a Twisted-Pair Cable 

4- 3 Ranges of Validity of Coplanarity Assumptions 

4- 4 Geometry of the Transmitting Loop and Transmission Line for the Case of Figure 4-l(a) 

4-5 Equivalent Transmission Line Circuit for - Induced Voltage and Current (Case of Zl = Z2 - 

zo) 

4- 6 Comparison of Loop-to-Loop-via-the-Line Signal 

Levels in Air with Equivalent Receiver Noise 

and Representative RMS Magnetic Field Noise 

Levels Measured in the McElroy Coal Mine 

4- 7 Comparison of Loop-to-Loop-via-the-Line Signal 

Levels in Air with Equivalent Receiver Noise 

and Representative RMS Magnetic Field Noise 

Levels Measured in the Robena and Itmann Coal 

Mines 

5- 1 Hybrid Noise Cancelling Diversity Receiving 

Technique (Simplified Basic Operation) 

5-2 Correlated Noise Vector Diagram 

Page 

77 

5- 3 Signal Factor Circle Diagram 

Arthur D Little, Inc. 
LIST OF FIGURES (Continued) 

Figure No. Page 

5-4 Signal Factor of I versus Phase Angle 8 in the 

Presence of an E-Field Signal SE 134 

5-5 Circle Diagram for Signal and Noise Factors of I 136 

5-6 Circle Diagram Indicating Region where I -< 1 136 

5-7 Expected Value of the Improvement Factor <I> in 

dB versus l/a and (REC/RH~) for Two Values of 

$/a (0.1, 1.0) and <8> = 1~12 139 

5- 8 Correction Term Q to <I> versus Phase Angle 8 

and Ratio ~~~1%~ 140 

Boundary Plot of R~~/R~~ versus l/a for Specified 

Improvement Factors <I> with f3/a = 1 142 

xii - 

Arthur D Little Inc. 
I. EXECUTIVE SUMMARY 

A. OBJECTIVE 

The objective of this task was to study and evaluate the feasibility 

of developing compact transmit antennas and/or other means for effici.ently 

coupling VLF to MI? radio energy between portable wireless communication 

units in coal mines. The Bureau of Mines would like to achieve a 

portable-to-portable range of about 1350 ft., especially in conductorfree areas of typical U. S. room and pillar coal mines, namely areas 

whereelectrical conductors such as power cables, trolley lines, and 

communication lines are absent. In the presence of these conductors, 

achieving the desired communication range is usually not a problem. The 

1350 ft. conductor-free range corresponds to that required for two miners 

to communicate between the leftmost and rightmost entries (spaced about 

600 ft. apart) of a seven-entry mine development while also being 

separated by a longitudinal distance of about 1000 ft. along the entries. 

This requires an area-coverage capability that is not adversely affected 

by the presence of the largely rectilinear grid of tunnels in the coal 

seam. Thus, the frequency band from 30 kHz to 1 MHz was selected as 

the band of primary interest for this investigation. 

B. APPROACH 

First, the ranges for completely wireless communication between 

two portable radios using practically-sized loop antennas in representative coal mine environments were estimated. Antenna technology in 

the VLF to MF band was then assessed in light of the range, intrinsic 

safety, and wearability requirements, to determine the most suitable 

antenna types for the application, and whether any of them could 

significantly improve upon the overall performance predicted for the 

reference loop antenna used in making the range estimates. Effort was 

also directed in two related areas. The first was a study to determine 

whether inductive coupling to cables and other conducting structures 

in the mine could substantially increase the communications range, since 

such cables, etc., are usually present where men are working underground. 

Arthur D Little, Inc 
The second was a study to assess the potential benefits of using a diversity reception noise cancelling technique, that utilizes both the electric and magnetic field components of the ambient noise, to substantially 

improve the output signal-to-noise ratio of the receiver when a portable 

unit must be operated in the vicinity of a strong source of electrical 

noise. 

C . FINDINGS AND CONCLUSIONS 

1. Wireless Range Estimates 

Estimates were made of the wireless communication range attainable 

in an underground coal mine in the presence of mine generated electromagnetic noise, for man-carried radios of practical size and source 

strength operating in the frequency band from 10 kHz to 1 MHz. A 

conventional air-core bandolier-type loop antenna of strength M = 0.5 to 

2 0.7 ampere-m was used as a reference source to gage the amount of improvement in antenna performance required, if any deficiency in range, and 

therefore signal strength, must be overcome solely through improved 

antenna design. 

It is shown that for mines having an effective conductivity of 

10-*Mho/m (near the upper limit for coal), the deficiency in signal-tonoise ratio, and thus range, is insuperable with man-carried equipment 

at any frequency in the band of interest. In mines for which a threelayer propagation model is applicable and the effective conductivity of 

-4 the coal is about 10 Mho/m (near the lower limit for coal), the deficiencies become much more manageable, particularly at frequencies between 

about 300 kHz and 1 MHz. Here, moderate improvements in antenna/ 

transmitter design and/or reception techniques may be able to provide 

modest increases in comunication range, and reduct ions in battery size 

and therefore weight of the portable units. However, the most likely 

value or range of values for the in-situ conductivity of U. S. bituminous 

coal must still be determined before a general conclusion can be drawn 

on the percentage of mines in which the desired range goal in completely 

conductor-free areas is likely to be achievable. 

I. 

Arthur D Little lnc. 
2. Antenna Technology 

Transmit antenna technology available in the VLF through MF bands 

was examined for its utility in portable mine wireless radio applications. In particular, an assessment was made of the feasibility of 

developing compact, portable transmit antennas that will efficiently 

generate radio waves in coal mines. The size of such antennas relative 

to wavelength classifies them as electrically small antennas which, by 

their very nature, are poor radiators. No major breakthroughs have 

occurred, or are likely to occur, to change this fact. Thus, the VLFMF mine wireless communication problem is one of optimizing the near 

field and induction field coupling between two loosely coupled portable 

electromagnetic field transducers (antennas) in the physical and noise 

environment of a mine. The problem is made more difficult by the mine's 

lossy conducting medium which introduces considerable signal attenuation. 

As a result, the choice of a specific antenna should not be based 

on its radiation efficiency. Instead it should be based on the overall 

power efficiency and practicality achievable by the complete transmitterantenna system in producing the largest usable signal at the desired 

range within the practical constraints of system overall size, werght , 

convenience of use, intrinsic safety, and ruggedness for roving miners. 

Thus, it is concluded that conventional air-core bandolier loop 

antennas, and perhaps asmall ferrite-loaded loop antennas, will be the 

most suitable and reasonable choices for roving miner portable radio 

applications at frequencies below about 1 MHz. Furthermore, a transmit 

2 

moment of about 2.5 amp-m (peak) appears to be a practical upper bound 

for intrinsically safe portable units for use by miners. Therefore, the 

completely wireless range goal of 1350 feet will probably be achievable 

only in the most favorable coal mine propagationand noise environments, 

and range capabilities will not be significantly better than those 

2 predicted for units having a transmit moment of 0.7 amp-m . For fixed 

station applications, horizontal-wire and vertical-rod mode exciters 

may also offer comparable or better performance than planar air-core 

loops, but in-mine measurements will be needed to resolve this matter. 

Arthur D Little, Jnc. 
1 MEz band when the loops are within about 2 meters of unloaded 

.: trolley wire/rails ; 

coupling to twisted pair cables is severely reduced over that 

for untwisted pairs; 

performance should be significantly reduced for loops located 

in tunnels adjacent to the one in which the conductors are 

located. 

4. Noise Cancelling Diversity Reception 

A hybrid noise cancelling diversity receiving technique was 

examined that would try to capitalize on the high levels of both 

electric and magnetic field noise components present in many parts of 

a mine at the frequencies of interest below 1 MHz. It is shown that 

if the instantaneous correlation is high enough between the noise E and 

H field components, it should be possible to improve the output signalto-noise performance of a mine wireless system by cancelling a major 

portion of the H-field noise picked up in a loop antenna by properly 

m 

Arthur D Little Inc. 
combining it (manually or automatically) with the E-field noise picked 

up by a small dipole or whip antenna that would be insensitive to H-field 

signals and the ambient H-field noise. This selectivity between signal 

and noise is likely for portable units using loop antennas that generate 

primarily H fields which couple to the receiver of other portables 

primarily through magnetic induct ion. 

Calculations to date indicate that potential signal-to-noise 

improvements between 10-15 dB may be possible when the E and H noise 

components are 80% to 90% correlated, a condition which may prevail in 

the vicintiy of severe noise sources. Such a S/N increase could then 

result in extending the range of a wireless communications system, or 

reducing the power required to attain a given range in the presence of 

high ambient electrical noise near one of the receiving units. Some 

basic field measurements are required to determine whether the degree 

of correlation between the E and H noise field components is high 

enough to warrant a more rigorous analysis and further development 

effort. 

Arthur D Little, Inc 
(This page intentionally left blank) 

- 

Arthur D Little lnc 
11. MINE WIRELESS RADIO RANGE ESTIMATES 

A. SUMMARY 

The feasibility of achieving a wireless communication range of 

1,350 ft (412 m) in an underground coal mine, in the presence of mine 

generated electromagnetic noise, with man-carried radios of practical 

size and source strength in the frequency band from 10 kHz to 1MHz is 

assessed. A conventional air-core bandolier-type loop antenna of strength 

2 M = 0.5 to 0.7 ampere-m is used as a reference source, to gage the 

amount of improvement required if any deficiency in range,and therefore 

signal strength,must be overcome solely through improved antenna design. 

-2 It is shown that for mines having an effective conductivity of 10 mo/m 

(near the upper limit for coal), the deficiency in signal-to-noise ratio, 

and thus range, is insuperable (by a vast margin) with man-carried equipment at any frequency in the band of interest. In mines for which a threelayer propagation model is applicable and the effective conductivity of 

the coal. is about loo4 Mho/m (near the lower limit for coal) , the def iciencies become much more manageable, particularly at frequencies between 

about 300 kHz and 1 MHz. Here, moderate improvements in antennaltransmitter design and/or reception techniques may be able to provide practical 

solutions. However, the most likely value or range of values for the 

in-situ conductivity of U.S. bituminous coal must still be determined 

before a generally applicable system can be developed for achieving the 

desired range goal in completely conductor-free areas of mines. 

B . INTRODUCTION 

The objective of this chapter is to establish some reference levels 

of performance that can be expected in a mine environment. A welldefined, practical sized antenna that can be conveniently worn on the 

body of a mobile miner, namely an air-core, multi-turn bandolier-type 

loop having a transmit magnetic moment in the range of 0.5 to 2 ampere2 

meter , is used as a reference signal source. The theoretical field 

strength generated by this source over the 1,350 ft range of interest 

is calculated for two electromagnetic wave propagation models for the 

underground coal mine. These signal levels are then compared with 

Arthur D Little, lnc. 
tunnels in the coal seam and the air-earth interface above the mine is 

ignored, reasonable simplifying assumptions for the frequencies, mine 

tunnel cross-sections, and mine depths of interest. In each instance we 

compute the value of the magnitude of the component of magnetic field 

perpendicular to the plane of the loop, for observation points which lie 

on a radial line in the plane of the loop. We designate this component 

asIHZlin the following equations. For convenient reference and comparison 

with this conducting medium case, we have also computed values for the 

same magnetic field component ~roduced by the same loop when placed in 

an a ir medium. 

1. Signal Field Strengths 

If r is the distance of the observation point fromthe loop, f 

is the frequency of operation, and M is the magnetic dipole moment, then 

the magnitude of the field componentl~ I as a function of distance and 

z 

frequency is given for both the air and conducting medium cases by the 

conventional field equations below (with terms grouped as shown for 

convenience). 

I 

Arthur D Little Inc. 
Air Medium 

where 

and c = (eOIIO)-' is the velocity of light in free space (3) 

Conducting Medium 

where C = r(oPoo)'= fir16 (5) 

and o is the electrical conductivity of the medium 

6 is the skin depth 

The conducting medium equations are those that apply when displacement 

currents are negligible, a reasonable approximation in a coal/rock medium 

of assumed conductivity a = ~ho/m at operating frequencies below 

a few MHz. 

Graphs of the field strength magnitudes are plotted versus frequency, 

with range from the loop as a parameter, in Figures 2-1 and 2-2 for air 

-2 and conducting media respectively. A nominal value of a = 10 Mho/m 

was used for the conducting medium (near the upper limit for coal). In 

both cases a source magnetic dipole moment M=NIA of value 0.5 ampere-meter 

2 

was used, a conservative practical value for a man-carried radio. The 

plotted field strengths may be interpreted as peak or rms amplitudes 

accordingly as M is either the peak or rms dipole moment. Note that for 

the air medium, Figure 2-1 shows that at given distance the magnetic 

field 1 H~ 1 eventually increases with increasing frequency. This occurs 

as the radiation field term exceeds the induction field terms, thereby 

leading to higher field strengths at higher frequencies. Figure 2-2 shows 

that the converse is true in a conducting medium. Namely for the 

-2 

a = 10 Mho/m case, as the frequency is increased, the increasing 

radiation field term is more than overpowered by an exponential attenuation factor caused by the resistive losses in the medium. Thus, the 

maximum achievable ranges are severely reduced in the conducting medium 

over that possible in air, and it is seen that in this conducting medium 

higher field strengths are obtained by reducing the operating frequency 

instead of raising it. 

Arthur D Little, lnc 
lalaw lad dwe 1 al gp u! lZHl 

10 

Arthur D Little, Inc 
-250 a 

a 

E 

lu 

I-'Y-' 

? 

&---- 

- *-- ------ -- - 

-4. 

r = 30 

- 

- M 

- 

M = 0.5 amp turn m 

- 'coal = 1 o'~ mhoslm. 

r = distance from signal loop in meters 

- - Signal Field 

- Noise Fields (measured at spot frequencies 

with 1 kHz bandwidth instrumentation) 

A,- Robena Mine No. 4, Figs. 49-53, Ref. 1 - 

em-,,, McElroy Mine, Figs. 4-28 to 4-35, Ref. 2 

- ltmann Mine No. 3, Figs. 9e, 10e, 1 le, Ref. 3 

- 

Frequency, kHz 

Source: Arthur D. Little, Inc. and National Buerau of Standards. 

FIGURE 2-2 MAGNETIC FIELD STRENGTH /Hz/ FROM A LOOP ANTENNA IN A CONDUCTING MEDIU

AND RMS MAGNETIC FIELD NOISE IN MINES (NORMALIZED TO 2.5kHz BANDWIDTH) 
2. Range Estimates 

a. Single Sideband Voice 

Plotted in Figure 2-2 are magnetic field rms noise levels measured 

in three coal mines(', 3, for comparison with the signal strength 

curves. The noise levels have been normalized for a single sideband AM 

voice system effective bandwidth of 2.5 kHz. These noise data were all 

taken in mine working sections with instrumentation having an effective 

bandwidth of 1 kHz at spot frequencies. At the Robena and McElroy mines 

the data were taken about 100 meters from working faces. The dip in 

noise level around 130 kHz at Robena was concluded to be atypically low 

and should not be considered representative. Both Robena and McElroy 

mines employ room and pillar continuous mining techniques. At the Itmann 

No. 3 mine, data were taken near the face area of a longwall working 

section. 

In Figure 2-3, the signal levels of Figure 2-2 have been replotted 

for comparison with samples of wideband tape recordings of magnetic 

(2) field n~ise levels measured near operating machinery in four mines . The noise levels have again been normalized to a bandwidth of 2.5 kHz. 

Figure 2-5 contains a similar comparison of signal and noise levels, but 

for noise level samples measured in rail haulageways. The noise levels 

in Figures 2-3 and 2-4 were obtained by narrowband FFT spectrum analysis 

of wideband tape recordings having upper cut off frequencies of 100 kHz 

(~obena) and 200 kHz (other mines). 

1. W. D. Bensema, M. Kanda and J. W. Adams, "Electromagnetic Noise 

in Robena No. 4 Coal ~ine", NBS Technical Note 654, April, 1974- 

2. M. Kanda, J. W. Adams and W. D. Bensema, "Electromagnetic Noise in 

McElroy Mine", NBSIR 74-389, June, 1974. 

3. M. Kanda, "Time and Amplitude Statistics for Electromagnetic Noise 

in Mines", NBSIR 74-378, June, 1974. 

Arthur D Little Inc. 
- Signal FieM 

r = distance from signal loop in meters 

M = 0.5 amp turn m2 

'coal = mhos/m. 

C 

"Spectra of Noise Fields in Mines, Ref. 2 

(measured with wideband tape recording 

instrumentation) 

- --- Noise Measurement System Self-Noise 

-10 - 

--I50 - 

-170 - 

-190 - 

--210 - 

0 20 40 60 80 100 120 140 160 1 80 200 

Frequency, kHz 

Source: Arthur D. Little, Inc. and National Bureau of Standards. 

FIGURE 2-3 COMPARISON OF LOOP SIGNAL IN CONDUCTING MEDIUM (FROM 

FIGURE 2-2) AND MINE NOISE NEAR OPERATING MACHINERY 

(NORMALIZED TO 2.5kHz BANDWIDTH) 

Arthur D Little Inc 
4 I 1 I 1 1 I I 1 1 1 

0 20 40 60 80 100 120 1 40 1 60 180 200 

Source: Arthur D. Little, Inc. and National Buerau of Standards 

Frequency, kHz 

FIGURE 2-4 COMPARISON OF LOOP SIGNAL IN C;ONDUCTING MEDIUM (FROM 

FIGURE 2-2) AND MINE NOISE IN HAULAGEWAY 

(NORMALIZED TO 2.5kHz BANDWIDTH) 

Arthur D Little. Inc. 
r = 30 

- 

- 

M = 1500 amp turn m 

'coal = 1 o-~ mhoslm. - 

r = distance from signal loop in meters 

- Signal Field 

- 

Noise Fields (measured at spot frequencies - with -- 1 kHz bandwidth instrumentation) 

A- - - Robena Mine No. 4, Ref. 1, Figs. 49-53 

- 

a- - - -- McElroy Mine, Ref. 2, Figs. 4-28 to 4-35 

rn ltmann Mine No. 3, Ref. 3, Figs. 9e, 10e, Ile - 

- 

Frequency, kHz 

Source: Arthur D. Little, Inc. and National Buerau of Standards. 

FIGURE 2-5 MAGNETIC FIELD STRENGTH /Hz/ FROM A LOOP ANTENNA IN A CONDUCTING MEDIUM 

AND RMS MAGNETIC FIELD NOISE IN MINES (NORMALIZED TO A 10Hz BANDWIDTH) 
Examination of Figures 2-2, 2-3 and 2-4 reveals achievable ranges 

in the vicinity of about 50 meters for marginal quality conventional SSB 

voice communication, and then only above about 100-200 kHz. These range 

expectations, based on direct-loop-to-loop coupling between man-sized 

-2 loops in a homogeneous conducting medium of 0 = 10 Mho/m.. (without the 

aid of any nearby conductors) in the presence of representative noise 

levels in operating mines, are unacceptably low compared with the 

desired range of 1,350 ft (412 m). At "quiet" times or locations these 

performance ranges can be expected to increase somewhat, but probably 

not drastically if the mine is not in an emergency power-down condition 

2 Increasing the transmit moment from 0.5 to 2 amp - m will not make a 

significant difference either. Section D of this chapter examines the 

range implications for a different propagation model which to date has 

been found to apply in several mines. 

b. Narrowband Paging 

A second reference example (a somewhat extreme one) has also been 

-2 examined for the same homogeneous conducting medium having a= 10 ~ho/m, 

to see if it would produce a significant improvement in range. Namely, 

it was assumed that the very large transmit moment (M = 1500 ampere - m 

2 

for a pillar - encircling single-turn loop) and the very narrow bandwidth (B = 10Hz) associated with recently developed audio frequency call 

alert mine paging systems could be safely generated and maintained over 

the 10 kHz - 1 MHz frequency band of interest. Such a transmitter 

strength and configuration might be associated with a fixed base station 

or repeater. The corresponding signal and noise field strength levels 

were computed and compared with each other to determine the increase in 

range that would occur. The above signal level and bandwidth changes 

represent a 94 dB improvement in signal-to-noise ratio. In practice 

such a large increase will be unattainable at the higher frequencies 

because of the high voltages required (because of the loop inductance) 

to generate the current levels in the transmit loop to produce the M of 

1500 amp-m2, and because the pillar-encircling loop lying in the horizontal plane is unsuitable for exciting the favorable TEM propagation mode 

in a three-layer model, which now appears to be a more represenative 

model for coal mines. 

16 I 

rthur D Little Inc. 
The signal field strengths were calculated as before, using the 

infinitesimal magnetic dipole field approximation, which will be accurate 

enough for estimating maximum operating ranges beyond 100 meters, in 

spite of the rather large finite size of the pillar-encircling loop. 

Thus, the signal field strength curves can be obtained by scaling up the 

Figure 2-2 signal curves, and the noise curves can be obtained by sealing down the Figure 2-2 noise curves. The rescaled signal and noise 

curves are plotted in Figure 2-5. These curves indicate that between 

about 200 kHz and 1 MHz, ranges would be limited to about 100-200 meters, 

but that between about 10 kHz and 200 kHz, it may be possible to achieve 

communication ranges out to between 200 and 300 meters, which still fall 

short of the 1,350 ft (412 m) range goal. Furthermore, this would 

represent only the "talk out" range from a base station. Because of the 

significantly lower signal strength available from a man-carried transmitter compared to that available from the pilar mounted transmitter, 

the "talk back" range from the mobile miner would still be substantially 

smaller and inadequate. 

Thus, we find that if the mine environment is found to behave like 

-2 

a homogeneous conducting medium having a conductivity of 10 Mho/m or 

greater, the desired range goal will be unattainable. However, it has 

been found that some mines may possess much more favorable wave propagation characteristics that tend to improve the outlook. These are 

discussed in the following section of this chapter. 

Arthur D Little Inc. 
D. THE THREE-LAYER MODEL 

A second, and more plausible model, namely the three-layer propagation model has been developed recently by Arthur D. Little, Inc. (4, 5) 

to explain the unexpectedly favorable results of radio transmission 

measurements(6) made at frequencies in the 50 kHz to 1 MHz band with 

loop antennas in a conductor-free area of a coal mine, Consolidation 

Coal's Ireland Mine near Moundsville, West Virginia. The measurements 

indicate that communication ranges in excess of 1,000 ft are attainable 

in this mine when both transmit and receive loop antennas are oriented 

to lie in a common vertical plane. The model which explains the observed 

behavior is one in which the coal is assumed to have a low conductivity 

compared with that of the surrounding rock above and below the seam. 

The propagation mode is then considered to be approximately a dipolar, 

two dimensional TEM mode with vertical E-field and circumferential Hfield in a horizontal slot (the coal seam) between two identical, higher 

conducting, half spaces (the rock). This model, as did the homogeneous 

medium model, ignores the presence of the mine tunnels, a good approximation in view of the large ratio of wavelength to tunnel dimensions. 

The model also represents a considerable simplification of the stratification of the layers of rock above and below the coal. We do not intend 

to justify this model or present detailed derivations here., since that is 

amply treated in the cited references. We only plan to excerpt, summarize, 

and use the results of that investigation for the purpose of assessing 

transmitter requirements to attain the 1,350 ft (412 m) communication 

range goal. 

4. Arthur D. Little, Inc., "Propagation of Radio Waves in Coal Mines" 

Chapter IV, Final Report on Task F, Task Order No. 1, Contract 

No. H0346045, October, 1975. 

5. A. G. Emslie and R. L. Lagace, "Propagation of Low and Medium 

Frequency Radio Waves in a Coal Seam", Radio Science, Vol. 11, 

No. 4, pp. 253-261, April, 1976. 

6. T. S. Cory, Summary Data Report No. 2 - "Mine Wireless Propagation 

Test Program - Ireland Mine Test Data at Medium Frequency", prepared 

for Collins Radio Group of Rockwell International for U.S. Bureau 

of Mines under Contract H0346067, Subcontract C-615171. 

- 

Arthur D Little Inc. 
1. Signal Field Strengths 

The three layer model is represented in Figure 2-6 which shows a 

cross section of the simplified geometry with the transmitting and 

receiving loop antennas in the vertical plane containing the path of 

propagation. The conductivity, a of the coal seam of thickness 2b is 

C' 

considered to be several orders of magnitude less than the conductivity 

a of the adjacent rock. The vertically-oriented transmitting loop r 

antenna produces, along the path of transmission in the coal seam, a 

horizontal magnetic field H and an approximately vertical electric @ 

field EZ. The fields are almost constant over the height of the coal 

seam. In the rock above and below the seam the fields die off exponentially in the positive and negative z-directions, respectively. At 

large radial distances from the antenna the fields decay exponentially 

at a rate determined by an effective attenuation constanta, which 

depends on losses both in the coal and in the rock and on the dielectric 

constant of the coal. There is also a 16 factor at large radial distances r 'due to the cylindrical spreading of the wave. 

The zero-order mode magnetic field H in the plane of the transmitting @ 

loop is given by 

where 

M = NIA is the magnetic moment of the transmitting loop antenna 

be = b + 112 6r is the effective half-height of the coal seam 

6r = Z-direction skin depth in the rock 

k = 6-iais the complex propagation constant in the radial direction r 

Hi2) ' (kr) = derivative of the first order Hankel function for an 

outgoing wave 

Arthur D Little Inc. 
Rock Or 

Approximate 

Field Directions 

h I --- -.

XMTR E~ 1 

t RCVS l-ct~i, 

// -7 2b Coal Seam 0 E 

c, c 1 1, ) 

I *. 

+ -.-- .- 

Rock 

or 

Source: Arthur D. Little, Inc. 

FIGURE 2-6 THREE LAYER MODEL GEOMETRY FOR LOW AND MEDIUM 

FREQUENCY RADIO WAVE PROPAGATION 

-80 

0 50 100 150 200 250 300 350 400 

Distance, (meters) 

Coal Seam Height 2b = 2 Meters - Loop Transmit Moment M = 0.7 amp -m2 

For oc= 1.4~ 1 o-~ Mholm 

Values of Coal Permittivity and Surrounding Rock Conductivity are 

Identical for all Curves, Namely Kc = 7, or = 1.0 Mholm 

Source: Arthur D. Little, I nc. 

FIGURE 2-7 THEORETICAL MAGNETIC FIELD STRENGTH PLOTS VERSUS DISTANCE AND 

FREQUENCY FOR THREE LAYER MODEL FOR TWO VALUES OF COAL 

CONDUCTIVITY, oc = 1.4 x 10-4 M~OI~ AND O, = 10-2 M~OI~ 111 

Arthur D Little Inc. 
A detailed derivation for , together with formulas for the phase and 

attenuation constants B and a expressed in terms of the conductivities 

(ac,ar ) of the coal and the rock, permittivity K of coal, the frequency C 

f and the half-height b of the coal seam are given in references 4 and 5. 

The common radial skin depth in both coal and rock is 6 = l/a . 

The best overall fit of the above propagation model to the Ireland 

mine experimental data was found (4, 5) to occur for coal and rock conductivities of 1.4 x loe4 Mho/m and 1.0 Mho/m, respectively, using an 

assumed coal dielectric constant of 7 (a value consistent with dielectric 

constant data obtained by NBS) C7)~lthough these values of conductivity 

are reasonable for bituminous coal and rock shales, they do lie close 

-4 

to extreme values for each. Namely, the value of 1.4 x 10 Mho/m for 

the conductivity a of the coal required to make the theory fit the 

C 

experimental data lies within the lower part of the range of conductivity 

values for bituminous coals; whereas the conductivity value 0 of 1.0 

r 

Mho/m required of the adjacent rock layers, although high, lies within 

the uppermost part of the range of reported conductivity values for some 

(8) types of shales and slates under certain conditions . Using these conductivity values, theoretical magnetic field strength 

curves have been computed and plotted in Figure 2-7 at three frequencies 

(57.5, 350, and 920 kHz) for a transmit loop source strength of M= 0.7 

ampere - m2 which is oriented and positioned as shown in Figure 2-6 in 

a high-coal seam of thickness 2b = 2 meters. Also plotted for comparison 

in Figure 2-7 are field strength curves at the three frequencies of 10, 

100, and 1000 kHz for the same source strength and configuration, seam 

thickness, and rock conductivity, but for the substantially higher coal 

7. D. A. Ellerbruch and J. W, Adams, "Microwave Measurement of Coal 

Layer ~hickness", Nat. Bureau Stand. (U.S.) NBSIR 74-387, 

September, 1974. 

8. E. I. Parkhomenko, Chapter 111, Electrical Resistivity of Rocks, 

Electrical Properties of Rocks, Translated from Russian and 

Edited by G. V. Keller, Plenum Press, N.Y. 1967 

Arthur D Little Inc. 
-2 conductivity of o = 10 Mho/m used in the homogeneous medium model in 

r 

Section C of this chapter. The higher value of oc lies within the 

upper part of the range of values for bituminous coal(8). Examination 

of Figure 2-7 reveals that dramatic changes in propagation loss can occur 

-4 

as the conductivity of the coal is increased from 10 to lom2 Mho/m. 

The low coal conductivity produces a substantially lower signal attenuation rate and only moderate variations of signal strength with frequency 

for the indicated ranges of distance and frequency. On the other hand, 

the higher coal conductivity results in substantially increased attenuation rates and large variations with frequency, that in turn cause the 

strongest signals to occur at the lowest frequencies, as in the homogeneous model. 

2. Range Estimates for FM Voice 

Plotted in Figures 2-8 and 2-9 are comprehensive noise plots which we 

have used together with the field strength curves of Figure 2-7 to arrive 

at new range estimates for wireless voice communication with man-carried 

personal radio systems in mines. We chose in this instance, a narrowband 

FM radio4 communication system having an IF bandwidth B = 12 kHz, receiver 

noise figure F = 6 dB, transmit magnetic moment M = 0.7 amp-m2,and loop 

2 effective turns-area NA = 1 m , similar to a system presently being developed for wireless mine communication applications by Collins Radio Group 

on Bureau of Mines Contract H0346047. The rms noise levels used to 

characterize the mine electromagnetic noise environment in Figures 2-8 

and 2-9 were derived from samples of magnetic field, time-averaged, 

rms noise levels measured (with 1 kHz bandwidth instrumentation) at 

spot frequencies in the frequency range of interest in three coal mines 

by Bensema, Kanda, and Adams 2' of the National Bureau of Standards. 

In the frequency band of interest, the average rms noise levels generally 

decrease at a rate on the order of l/f with increasing frequency, and 

exhibit relatively large variations during typical mine work shifts. The 

magnitude of these long term variations in average rms level also decrease 

with increasing frequency in the LF to MF band,commonly being on the order 

of 45-50 dB at frequencies below about 100 kHz and decreasing to about 

25-30 dB at frequencies around 1 MHz. 

9. W. D. Bensema, M. Kanda and J. W. Adams, "Electromagnetic Noise in 

Itmann Mine", Nat. Bur. Stand. (U.S.) NBSIR 74-390, June, 1974. 

Arthur D Little Inc 
Plotted Noise Levels Normalized to 12kHz Bandwidth - Measured with 1 kHz 

Instrumentation Bandwidth 

- - - Vertical Field Component 

- . - Vertical Component - "Quiet Time" 

Horizontal Field Component 

McElroy Mine - Continuous Miner Section and Nearby Rail Haulagewey 

o Near end of rail haulage line 

u Near intersection of rail haltlageway and conveyor belt 

4 Near operating continuous mining machine 

Near section power distribution center 

Frequency in kHz 

Source: National Bureau of Standards (Report NBSl R 74-389, June 1974) 

FIGURE 2-8 REPRESENTATIVE RMS MAGNETIC FIELD NOISE LEVELS 

MEASURED IN THE McELROY COAL MINE 

Arthur D Little. Inc 
Plotted Noise Levels Normalized to 12kHz Bandwidth - Measured with 1 kHz 

Instrumentation Bandwidth 

- - Vertical Field Component 

- - - Vertical Component - "Quiet Time" 

-Horizontal Field Component 

ltmann Mine - Longwall Panels 

4 At longwall face head end - Farley panel 

230 ft from longwall face - Farley panel 

n At longwall face head end near main conveyor belt - Cabin Creek panel 

Robena Mine - Rail Haulageway serving continuous miner section. All curves for 

same location approximately 300 meters from face area 

C) Horizontal (E-W) - 1st day 

+F Horizontal (N-S) - 1st day 

Q Vertical - 1st day 

'ti Vertical - 2ndday 

t Note: NBS be1ie;es that these three data 

points at 130kHz on the 1st day at 

Robena were atypically low and 

therefore should not be considered 

Frequency in kHz 

-50 

-60 

-70 

Source: National Bureau of Standards (Reports NBS Technical Note 654, April 1974 Robena; 

and NBSlR 74-390, June 1974, Itmann) 

- representative. 

- 

i I I 1 I I I I 1 1 I 

FIGURE 2-9 REPRESENTATIVE RMS MAGNETIC FIELD NOISE LEVELS 

MEASURED IN THE ROBENA AND ITMANN COAL MINES 

I* 

Arthur D Little. Inc. 
The wireless communication ranges are estimated for the above radio 

system and noise parameters and a radio circuit grade of performance goal 

within the Circuit Merit Figure /I3 classification, the minimum figure 

normally considered for commercial radio service. To obtain this level 

of performance the minimum average rms carrier-to-noise ratio at the 

receiver must be at least 10 dB or better for the mine noise conditions 

prevailing during the communication intervals. 

The range estimates are shown in Table 2-1 for active-mine electromagnetic noise conditions and quiet receiver-noise-limited conditions. 

Table 2-1 shows that only very short and inadequate communication ranges 

are predicted for all noise conditions and frequencies for the high con- 

-2 ductivity coal case (a = 1.0 x 10 Mho/m), as was the case for the 

C 

homogeneous medium model. The extremely high rates of signal attenuation 

for this case also make communication ranges very insensitive to even 

large upward or downward changes in local levels of average rms electro- 

-4 magnetic noise. For the low conductivity coal case ( u = 1.4 x 10 Mho/m), C 

the communication ranges are substantially increased to more practical 

values that approach the 412 m (1,350 ft) range goal. However, these 

extended ranges are also more sensitive to variations in local rms 

electromagnetic noise levels, because of the greatly reduced rates of 

signal attenuation when the coal conductivity is low. This effect is 

illustrated in Table 2-1 by the wide spans in estimated ranges under 

active-mine noise conditions. Active-mine noise levels can also 

occasionally fall below intrinsic receiver noise levels. When this 

occurs, the maximum range will be limited by, and identical to, that 

dictated by receiver noise not mine noise. These receiver-noise-limited 

ranges are also shown in Table 2-1. 

Arthur D Little, Inc. 
TABLE 2-1 

RANGE ESTIMATES FOR WIRELESS COMMUNICATION WITH PERSONAL PORTABLE FM 

RADIOS IN A HIGH-COAL SEAM 

(Seam Height of 2m, and Kc = 7, bounded by rock of conductivity o = r -2 1.0 Mho/m, for coal conductivities of 1.0 x 10 and 1.4 x Mho/m) 

Frequency 

(kHz) 

Coal Conductivity 

0 (Mho/m) C 

Source: Arthur D. Little, Inc. 

Communication Range in Meters(ft) 

Noise Conditions 

Active 

Receiver Noise Mine Noise 

- 

Arthur D Little lnc. 
3. Discussion of Implications 

The findings of the three layer model based on the Ireland mine 

experimental data indicate that communication ranges within striking 

distance of the 1,350 ft (412 m) goal are attainable under some noise 

conditions in at least one mine having the right combination of rock 

and coal conductivities, in particular, a low value of coal conductivity. 

If this behavior is found to be common in coal mines, it is conceivable 

that the range goal could be reached through the clever application of 

of currently available antenna and transmitter technology. However, 

it is reported '8 t h at the conductivity of coal can take on a wide range of 

values, encompassing at least two orders of magnitude, which could 

result in different seams or mines within the same seam having drastically 

different attainable communication ranges. Therefore, it is important 

to determine whether the favorable Ireland mine conditions are typical 

or exceptional for bituminous coal mines in the United States. This 

should be accomplished by performing similar propagation measurements 

in several more coal mines in several of the major coal seams, to obtain 

data from mines having both similar and different geological properties 

regarding both the coal seam and the surrounding rock. Only then will 

it be possible to confidently determine the most favorable operating 

frequency, and the associated performance limits, for a mine wireless 

communication system using currently available technology in the 10 kHz 

to 1 MHz band. 

Such a measurement program has been defined and is presently being 

implemented for the Bureau of Mines by a measurement and analysis team 

consisting of T. Cory/Spectra Associates/Collins Radio Group on 

Contract H0366028 and Arthur D. Little, Inc., on Contract H0346045, Task 

Order No. 4, respectively. The preliminary findings to date indicate 

that two other mines also in northern West Virginia and inthe Pittsburgh 

seam exhibit behavior and coal conductivities similar to that found in 

the Ireland mine, while another mine in Illinois in the Herrin No. 6 

seam exhibits significantly higher signal attenuation rates by about a 

-3 factor of three and a coal conductivity of about 10 Mho/m. Thus, the 

issue remains unresolved at the time of this writing. However, if large 

Arthur D Little, lnc 
variations in conductivity turn out to be common from mine-to-mine, it 

is unlikely that improved antenna technology alone will ever be capable 

of providing the significant improvements in system performance that will 

be required for 10 kHz to 1 MHz operation in mines with high conductivity coal. An assessment of the likely impact of antenna technology 

on performance is discussed in the following chapter. 

It should be noted that the quiet-area range predictions of 

Table 2-1 should be most applicable for conductor free areas such as 

the part of the Ireland mine in which the signal propagation measurements 

were made; whereas the active-area range predicitons are probably pessimistic, because the active, noisy areas are generally those where conductors such as trolley wire/rails and power and telephone cables are 

also present. In the latter situation one would expect that communication ranges could be considerably increased by the lower-loss transmission 

line type of propagation made possible by the inductive coupling of the 

fields from man-carried loops to such nearby conductors. Single conductors or cables will act as one element of a transmission line, with 

the highly conducting rock and/or moderately conducting coal serving as 

the return current path, while trolley lines have the rails as return 

conductors. Since miners requiring radio communications are very likely 

to be working in the vicinity of such conducting structures, this situation, as opposed to the completely conductor-free situation, may become 

an equally important operational requirement influencing the choice of 

frequency. Therefore, the coupling of loop antennas to two-wire transmission line structures is treated in Chapter IV. On the other hand, 

methods of reducing the deleterious effects of mine generated radio 

noise should not be overlooked as a means of improving the signal-tonoise performance and thus the communication range. For example, the 

effective level of received noise can in some cases be reduced by up to 

10-20 dB by simple processing, such as clipping and blanking, if the 

noise is highly impulsive in nature. The potential benefits of yet 

another method, a special noise cancelling diversity reception technique 

that may be particularly suited to the mine noise environment, are presented in Chapter V. 

Arthur D Little Inc. 
111. ANTENNA TECHNOLOGY 

A. SUMMARY 

Transmit antenna technology available in the VLF through MF bands 

is examined for its utility in portable mine wireless radio applications. 

In particular, an assessment is made of the feasibility of developing 

compact, portable transmit antennas that will efficiently generate 

radio waves in coal mines. The size of such antennas relative to wavelength classifies them as electrically small antennas which, by their 

very nature, are poor radiators. No major breakthroughs have occurred, 

or are likely to occur, to change this fact. Thus, the VLF-MF mine 

wireless communication prcblem is one of optimizing the near field and 

induction field coupling between two loosely coupled portable electromagnetic field transducers (antennas) in the physical and no<se environment of a mine. The problem is made more difficult by the mine's lossy 

conducting medium which introduces considerable signal attenuation. 

The choice of a specific antenna should, therefore, not be based 

on its radiation efficiency. Instead it should be based on the overall 

power efficiency and practicality achievable by the complete transmitterantenna system, in producing the largest usable signal at the desired 

range within the practical constraints of system overall size, weight, 

convenience of use, intrinsic safety, and ruggedness for roving miners. 

Thus, it is concluded that conventional air-core bandolier loop antennas, 

and perhaps smaller ferrite-loaded loop antennas, will continue to be 

the most suitable and reasonable choice for roving miner portable radio 

applications at frequencies below about 1 MHz. Furthermore, a transmit 

moment of about 2.5 amp-m2 (peak) appears to be a practical upper bound 

for intrinsically safe portable units for use by miners. Thus, the 

completely wireless range capabilities will not be significantly better 

2 than those predicted for units having a transmit moment of 0.7 amp-m . For fixed station applications, horizontal-wire and vertical-rod mode 

exciters may also offer comparable or better performance than planar 

air-core loops, but in-mine measurements will be needed to resolve the 

matter. 

Arthur D Little, lnc. 
The following sections of this chapter treat some of the basic 

principles and limitations of small antennas operating below 1 MHz in 

both free space and conducting media, together with an overview of 

specific antenna types considered for manpack portable applications and 

for base station applications. The types of antennas treated are the 

class of electrically small antennas in general and, specifically, 

conventional whip antennas, active whip antennas, conventional air-core 

loop bandolier antennas, ferrite loaded loops, specially resonated multiturn air core loops, self-resonant air-core helical loop antennas, 

horizontal-wire mode exciters, and vertical-rod mode exciters. There 

is an extensive body of published reference information which covers 

the subject matter in considerable detail. The approach taken in this 

chapter is to provide an overview of this antenna technology and its 

significance to the mine communications problem at hand. The reader is 

referred to the specific references to obtain a more detailed treatment 

of the material. 

Arthur D Little. lnc 
B. ELECTRICALLY SMALL ANTENNAS 

The desire is to obtain a small, lightweight, unobtrusive, effective 

antenna that can be conveniently worn by a roving miner for use with a 

personal radio operating somewhere in the VLF to MF range of 10 kHz to 

1000 kHz (1 MHz). The antenna's impedance characteristics should also 

be relatively unaffected by the antenna's position on the miner's body 

and the antenna's proximity to his body or structures found in underground coal mines. 

Resonant length dipoles, monopoles, and loops have maximum dimensions 

on the order of a quarter to one-half wavelengths, which make them good 

radiators, namely, good couplers of power from a transmitter to the 

surrounding space. They produce radiation resistances typically on the 

order of 35 to 80 ohms and zero net energy storage, or reactive impedance, 

at their resonant length frequencies. Therefore, they can achieve very 

high efficiencies, be conveniently driven from conventional sources, and 

easily meet conventional single-channel bandwidth requirements. 

Electrically small antennas present a considerably different and 

more difficult design problem than the more conventional type of resonant 

length antennas. An electrically small antenna is defined as one whose 

size is a small fraction of the wavelength, more specifically less than 

1/10 of a wavelength. Such antennas possess fundamental limitations on 

their electrical characteristics and performance that are directly 

attributable to their size compared to wavelength alone. Portable 

antennas for roving miner applications at frequencies below 1 MHz fall 

well within the small antenna size limits. Therefore, their performance 

will in fact be governed by the limitations and characteristics imposed 

on electrically small antennas. 

1. Electrical Smallness 

Free space wavelengths for operating frequencies from 10 kHz to 

1000 kHz take on values of 30,000 to 300 meters respectively. Comparing 

these wavelengths to the maximum conceivable dimension d = 1 meter for 

a personal antenna worn by a roving miner, we get 1 <d< 1 --- 

30,000 X 300 ' 

Arthur D Little, Jnc. 
which immediately places such an antenna well inside the category of 

"electrically small" antennas, namely, antennas having d < - The 10' 

more the size of an antenna is decreased with respect to wavelength the 

more it behaves as a high-Q, lumped-element energy storage device and 

as a very poor radiator of power to the surrounding space. These 

limitations are fundamental and are imposed by the antenna's size 

compared to wavelength, as illustrated in a classic paper by Chu (1) in 

1948 and by ~arrin~ton'~) and Wheeler. (394) ~hus , electrically small, 

lossless, whip-type dipole antennas behave as capacitors, and loop-type 

antennas behave as inductors. Their input impedances are highly 

reactive and have only a very small resistive part defined as the 

radiation resistance--a resistance whose value decreases rapidly with 

decreasing frequency (i.e., increasing wavelength). 

2. Radiation Resistance 

The radiation resistance Rr is a measure of the power radiated by 

an antenna. It is given by (5) 

for dipole and loop antennas respectively, where h is the effective 

e 

height of the dipole and N and A are the number of turns and area of 

the loop. Note the strong inverse dependence on wavelength. The 

radiation resistance Rr is the real part of the input impedance of a 

12 lossless antenna, and -I Rr represents the real power radiated. 2 0 

Namely : 

where Pr is the total radiated power computed by integrating the 

electromngnetic power density in the far field over the surface of a sphere 

concentric with the antenna, and I. is the antenna input current which 

produces the radiated field. Substituting into (I), (2), and (3) 

practical antenna dimensions of0.5 meter diameter for a 10-turn 

loop, a 0 -25 meter effective height*for a dipole, and a 1 ampere 

input current to each from a portable mine wireless unit operating 

. at 300 kHz, we finif" that the radiation resistances and radiated powers 

------------------- 

*For an electrically small unloaded dipole having a triangular 

distribution of current which is maximum at the terminals and zero 

at the ends, the effective height is one-half the dipole length. 

- 

Arthur D Little, lnc. 
are: 

-6 R =5.5x10 ohm 

re 

-9 Rre = 1.5 x 10 ohm 

for the dipole and loop antenna respectively. At a frequency of 30 kHz, 

the upper boundary of the VLF band, the above values would of course be 

further reduced by a factor of - for the dipole and 100 for the 10,000 

loop. When compared with the resistive losses associated with practical 

antennas, these radiation resistances become insignificant. This fact 

accounts for the extremely low efficiencies attainable from man-carried 

antennas at these low operating frequencies. Thus, other factors take 

on a much greater importance than efficiency in forming a basis for 

antenna selection. 

3. Efficiencv 

The efficiency of the antenna alone is defined as the ratio of the 

power radiated to the sum of the power radiated and the power dissipated 

in the ohmic resistance Rd associated with the antenna conductors, 

This is the best that can be done. For example, if the above-mentioned 

10-turn, 0.5 meter diameter loop was made of /I16 AWG copper wire, its 

resistance of 0.73 ohm including skin and proximity effects would give 

the negligible antenna efficiency of only 2 x The efficiency 

of the transmitting system is further degraded by the resistance 

associated with the source imped'ance, and the resistance associated 

with any tuning circuitry that may be required to drive the highly 

reactive antenna from practical power sources. These equivalent 

resistances must be added to the value of R thereby reducing E . To d' a 

arrive at the overall system efficiency, which impacts on power supply 

requirements, and in particular battery life, account must also be 

taken of coupling coefficients and losses associated with impedance 

matching and coupling circuits,and the efficiency of the transmitter1 

power amplifier electronics too. See Wheeler (293) and Watt (6) for 

concise treatments of these topics. 

Arthur D Little lnc. 
4. Quality Factor-Q 

The high-Q, reactive nature of the small antenna points to the 

need for tuning and matching circuits to provide the transmitter with 

the proper real impedance load to optimize efficiency, power transfer, 

or some other performance parameter. The tuning and matching elements, 

like the antenna itself, are not ideal. Thus, resistive loss is introduced by these elements, which reduces the attainable efficiency and Q, 

and increases the antenna circuit bandwidth. 

Figure 3-1 from reference (7) presents a curve of the behavior of the 

intrinsic Q of electrically small antennas, based on the work of Chu. (1) 

The quantity "a" represents the radius of the minimum size sphere 

within which the antenna can be enclosed. This Q is a measure of the 

electromagnetic energy stored in the near field of the antenna compared 

to the power radiated into the far field by the antenna. Namely, 

- w x peak stored. ener 

'r average power radiattz ' 

Note the strong dependence on the radial dimension "a" expressed in 

wavelengths in Figure 3-1 and how rapidly Q increases with decreasing 

antenna size compared to wavelength, leading to the very narrow, 

intrinsic bandwidths characteristic of electrically small antennas. 

At an operating frequency of 300 kHz, the 0.5 meter maximum 

antenna size of interest gives a 27~a/X value of approximately 0.001, 

6 which from Figure 3-1 results in a Q well in excess of 10 . The 

r 

antenna's intrinsic radiation bandwidth (Af = fo/Q) is therefore less 

than about 0.1 Hz at 300 kHz. Fortunately, neither of these values of 

Qr or Af is desirable or attainable in practice for roving miner 

portable radio applications. For example, net Q's of about 100 are 

typical for the antenna and its tuning circuit as a result of resistive 

type losses. This results in a more practical bandwidth of 3 kHz at 

300 kHz, but in a poor net radiation efficiency of much less than 0.01% 

(from the alternate efficiency expression E = Q&,/Q~ presented below). 

5. Power Factor 

The intrinsic antenna behavior can also be conveniently expressed 

in terms of the radiation power factor introduced by Wheeler: (3) 

- 

Arthur D Little, lnc. 

where R is the radiation resistance and L and C are the intrinsic 

r 

inductance and capacitance of the loop and dipole antennas respectively. 

This concept of power factor is a convenient and descriptive one for 

understanding antenna behavior and limitations of performance. For 

example the expression for antenna efficiency, including the resistance 

of the tuning elements, can be written as 

where pf 1 

R = - wL = RRWC = - Q , R is the ohmic resistance of the antenna R 

and tuning elements, QR fs the Q due to these ohmic losses alone, and 

Qir is the Q which includes both ohmic and radiation losses. PfR will 

of course be increased further by the real part of the source impedance 

seen by the tuned antenna circuit, and pf will be decreased by the 

r 

efficiency of coupling to the tuned circuit. The reader is referred to 

Wheeler(3) for a more detailed presentation and for sample efficiency 

calculations for some specific antenna types. 

6. Effective Volume 

Wheeler also presents the extremely useful concepts of antenna 

effective volume, V', and the volume of what he denotes as a "radian 

sphere, It VS , or a "radian cube, " Vc. V' is the physical volume of the 

antenna multiplied by a shape factor greater than unity, ke for electric 

dipole-type antennas and k for magnetic dipole-type (loop) antennas, m 

as depicted in Figure 3-2 from Wheeler. (4) The shape factors are 

defined to have values such that the indicated inductance and capacitance 

formulas give the correct values of L and C for specific antenna 

geometries of interest. (394) V is the volume of a sphere of radius 

s 3 A equal to one radian length, namely - 27r ' and Vc is the volume of a cube 

with side dimensions of one radian length - It has been shown (1,8) 

2Tr0 

that the radiated power in the far field of an electrically small -1 A 

(d < antenna of either type is accompanied by stored energy which 

is mostly located in the antenna near field within the radian sphere. 

Yq 

Arthur D Little lnc 
Antenna 

TY pe L 

(Decreased By Dielectric Core) (Increased By Magnetic Core) 

Source: Reference 4. 

FIGURE 3-2 EFFECTIVE VOLUMES AND RADIATION POWER 

FACTORS FOR ELECTRICALLY SMALL ANTENNAS 

Arthur DLittleInc. 
Furthermore, the radiation power factor, pfr, which is the ratio of this 

radiated power to w times the stored energy, is found to be proportional 

to the ratio of the antenna's effective volume V' and the volume of 

either the radian sphere V or radian cube V . Namely, S C 

- 2 v' =-- 1 v' 

pfr -9y ' pfr 6r Vc (7,8) S 

~hese expressions allow a very convenient and rapid method of computing 

the radiation power factor (together with the Q and radiation resistance) 

and comparing it with the circuit ohmic loss power factor to determine 

the efficiency of the tuned antenna. 

Figure 3-3 illustrates how well the volumetric comparison method 

allows a quick grasp of the fundamental limitations imposed by antenna 

size alone on the radiation power factor, the Q, and the radiation resistance. Figure 3-4 depicts comparative examples (4) of the effective 

volumes of several common dipole and loop antenna types. The figure 

reveals that long and thin electric dipoles occupy a much larger 

effective volume than the volume of the physical conductors. Note 

also that the effective volume of a multiturn "flat" square loop is 

not much different from that of a corresponding single turn loop, and 

that the effective volume for both is approximately given by the 

volume of a sphere of diameter equal to the side dimension of the loops. 

It can also be shown (394) that short, flat air-core loops provide a 

much larger effective volume than long, thin, solenoid-type air-core 

loops of comparable maximum dimensions. The converse is true for 

electric dipole antennas; namely, the long thin type provides a better 

volume utilization than a short capacitor-plate type. 

An exception to this rule is the loop antenna loaded with a long 

thin high-permeability core as depicted in Figure 3-4. In this case 

the loop's effective volume is similar to that for a comparably-sized 

long thin dipole, where the loop dimensions now include those of the 

core; namely, the thin dipole antenna shape factor applies in computing 

the effective volume. Thus, in some applications a core-loaded loop 

may offer shape advantages that outweigh the somewhat greater 

Ilr 

Arthur D Little, Inc. 
Source: Arthur D. Little, Inc. 

FIGURE 3-3 ILLUSTRATION OF RADIATION POWER FACTOR LIMITATIONS 

IMPOSED BY ANTENNA EFFECTIVE VOLUME RELATIVE TO 

VOLUME OF A RADIAN SPHERE 

Arthur D Little, Inc. 
Wire Cone 

(a) Dipoles 

Discs 

Wire Strip 

(b) Loops 

Turns 

/With Ferrite Core Effective Volume 

- _With Air Core 

------ 

I 3, 

(c) Air And Ferrite Core Loops 

Source: Reference 4. 

FIGURE 3-4 COMPARATIVE EXAMPLES OF THE EFFECTIVE 

VOLUMES OF SEVERAL COMMON ANTENNA TYPES 

-8, 

Arthur D Little. Inc. 
effective volume offered by a flat air-core loop of equal maximum 

dimension. Additional limitations imposed by the properties of the 

core material must, of course, also be considered in practice. The 

opposite behavior is experienced by electric antennas loaded with 

dielectric material; namely, the effective volume is reduced by the 

presence of the material. 

Applying this effective volume method to either a thin electric 

dipole or flat air core loop of maximum dimensions d = 0.5 m gFves a 

9 radiation power factor and Q on the order of loW9 and 10 respectively. r 

Taking the ratio of Q to a practical tuning circuit Q of 100 

r Rr 

results in a negligible radiation efficiency of 0.00001%. 

7. Application Implications 

The above described fundamental limitations imposed on electrically 

small antennas are severe. Thus, it is not surprising that portable 

two-way radio systems for most conventional applications on the surface 

are designed to operate at frequencies in the VHF and UHF bands and to 

a lesser extent in the HF band, bands in which wavelengths become less 

than, or comparable to, the size of a person. However, the HF band, 

with the exception of some military applications, is still by and large 

mainly utilized for radio communication between either fixed stations 

or sizable mobile platforms such as ships, planes, and motorized 

vehicles,which can accommodate larger antennas than people can. Even 

man-carried HF military transceivers, such as the PRC-74, need to 

utilize clumsy inductively loaded whips about 2 meters long. (9) 

However, the efficiency of even optimally loaded 2 meter whips falls 

off rapidly at the low end of the HF band when the whip becomes less 

than about 0.05 A. (10) 

Below HF the problems get rapidly worse, even for fixed station 

transmitters. To achieve reasonable efficiencies, the ability to 

construct large, low loss antenna structures tuned by large low loss 

circuit elements becomes important. The tall radio towers used by 

AM broadcast stations in the MF band are perhaps some of the more 

familiar examples. (I1) At LF the problem worsens and the antenna 

structures and tuning circuits get larger and more difficult to 

Arthur D Little, Inc 
design, (11y12) such as the one shown in Figure 3-5. At VLF the antennas 

become huge structkes which stretch the state-of-the-art of both 

electrical and structural design technologies in order to achieve high 

efficiencies. l1 ' l3 Thus, VLF transmissions between fixed and 

mobile stations are generally one way, by necessity. 

Two Navy VLF transmitters, one at Cutler, Maine, (6) and the other 

at Northwest Cape, Australia, (694) represent extreme examples in that 

they are the largest antennas in the world and yet still fall within 

the defined limits of electrically small antennas. Figure 3-6 depicts 

plan and elevation drawings of the Australian antenna installation, (4) 

which is over 1000 feet high, more than 1.5 miles in diameter, and 

operates at a frequency of about 15 kHz (A = 20 km) at an efficiency 

greater than 50% (the Cutler, Maine, installation is greater than 80% 

efficient). It is basically a vertical electric monopole (whip) 

antenna over an extensive man-made ground plane. It is heavily toploaded with diamond-shaped cable grids to increase the antenna 

capacitance and provide a uniform vertical current distribution. These 

antennas are good examples of the extreme measures required at VLF to 

get high efficiency operation. For a comprehensive treatment of VLF 

radio systems in general and antenna system design problems in particular, 

the reader is referred to the text by Watt. (6) 

To attain the Navy's overall system objective of very long (globaltype) ranges, which at VLF takes a megawatt of radiated power, the 

option of throwing away high multiples of this radiated power into an 

inefficient antenna system is not available to the designers. As 

shown in Section III.B.3of this chapter, high efficiency requires the 

radiation resistance to be on the order of, and preferably greater than, 

all ohmic losses in the antenna structure and tuning and matching 

networks. To accomplish this at VLF requires as great a physical 

height as practically possible together with extensive top loading to 

increase the antenna's effective height, and thus the radiation resistance. 

It also requires the difficult and specialized design of the antenna 

structure, tuning elements, and ground plane (6 '11'12) to reduce all 

'possible ohmic losses to on the order of 0.1 ohm, the approximate value 

of the radiation resistance. 

H 

Arthur D Little, Inc. 
", 

Sour@: Reference 12. 
(a) Plan View 

f = 15.5 kHz h = 19.3 Km 

R, = 0.144 ohm X = 63 ohms 

P, = 1 MW VI = 435 MVA 

pf, = .0023 

(b) Elevation View 

Source: Reference 4. 

FIGURE 3-6 EXAMPLE OF STATE-OF-THE-ART 

LARGE VLF ANTENNA IN NORTHWEST CAPE AUSTRALIA 

LI 

Arthur D Little, Inc 
Since the antenna is still small electrically, it still behaves 

basically as an energy storage element having high reactance (63 ohm) 

compared to its radiation resistance (0.14 ohm). It therefore has a 

high intrinsic Q and narrow intrinsic bandwidth. One of the prices of 

achieving a high efficiency, by keeping the total ohmic resistance 

comparable to the low radiation resistance, is the antenna's narrow 

resonance bandwidth of about 130 Hz at 15 kHz. This is accompanied by 

huge amounts of reactive power, 435 MVA, 2.63 KA at 165 KV, which the 

antenna and tuning element structures must also accommodate, in addition 

to the real input power of 2 MW into about 0.3 ohm, 1 MW of which gets 

radiated. 

The narrow bandwidths imposed by high efficiency VLF antenna 

systems constrain such systems to coded data-type communications instead 

of voice. In the Navy VLF submarine communications application this 

affects the maximum allowable bit rate of transmission,which is not 

considered an unacceptable limitation. In a mine wireless radio voice 

communications application the high-Q, narrow bandwidth problem is a 

severe one, and of course unacceptable. The mine wireless application 

requires at leaet about 2.5 kHz of bandwidth for a single-sideband 

system and about 12 kHz for a conventional narrowband FM system. Thus, 

the transmit antenna size constraint for portable mine wireless voice 

communication applications, together with the system bandwidth required, 

make a goal of high radiation efficiency not only unattainable, but 

also undesirable for the VLF, LF, and even MF, radio bands from 10 kHz 

to 1000 kHz. 

8. Impact of Conducting Media 

When electrically small antennas are immersed in a conducting 

medium as in the case of the mine wireless application, additional 

factors come into play which dominate the choice of methods and 

antennas to optimize radio system performance. First and foremost is 

the severe signal attenuation introduced by the resistive losses 

experienced by the waves in the conducting medium. The conductivity 

of the medium introduces power dissipation external to the antenna 

conductors. This reduces both the radiated and reactive fields and 

Arthur D Little, Inc 
powers that would be present in a free space medium, thereby greatly 

reducing the attainable communication range. This effect is well depicted 

in Figure 2-1 in the preceding chapter. Furthermore, the free space 

definitions of radiation resistance and radiated power are no longer 

meaningful measures of performance or antenna efficiency. The so-called 

radiated power is no longer a fixed value independent of distance in the 

antenna far field, and the radiation resistance is no longer a measure 

of the input power coupled to or consumed by the medium external to the 

ant enna . 

Limitations imposed by signal attenuation and representative 

mine development practices have led to the relatively modest range 

goal of 1350 feet for the portable, or mine wireless system, a range 

which falls well within the free space near field region, and by and 

large, within the induction field region in a coal mine conducting 

medium. Consequently, the mine wireless cormnunication situation is 

reduced to a near and induction field coupling problem between two 

loosely coupled and mobile electromagnetic field transducers (antennas) 

embedded in a lossy medium. Thus, the choice of antenna for the 

portable mine wireless application will not be based on its radiation 

efficiency, but on: 

the overall power utilization efficiency of the transmitterantenna system in producing the largest usable signal strength 

at the desired range, and 

a the overall practicality of its implementation with regard to 

overall system size, weight, convenience, and mine worthiness 

for the roving miner application. 

Before proceeding to the discussion of specific antenna types in 

the following section, we compare the behavior of the received signals 

produced by the principal field components of electrically small loop 

and dipole transmit antennas embedded in a conducting medium. The 

portable antennas retain their electrically small classification because 

the reduced wavelength (defined as X = 2rx skin depth in the conducting 

medium) remains much greater than the antenna size for the frequencies 

and conductivities of interest. Equations for the Ee and He field 

Arthur D Little, Inc. 
components produced by ideal infinitesimal electric and magnetic dipole 

sources, respectively, are used to compare the loop-to-loop response 

with the dipole-to-dipole response. The field component equations apply 

for the condition a >> we, which generally applies throughout the 

frequency band and range of conductivities of interest for the portable 

mine wireless application. 

For dipoles : Ee - - Ih (1 + yr + ~~r~)e-~~sine 3 4rrr a 

NIA 22 - For loops : He = -(1 + yr + y r )e Yrsine 

4rrr 3 

2 2 where y = jwpa-u ep :: jwpa. 

The equations are for z-axis oriented infinitesimal dipole and loop 

antenna elements, (I4) and I is the uniform current in the antenna 

elements. The equations apply for a loop fully insulated from the 

medium, and a dipole insulated along its length but grounded to the 

medium at each end of the dipole. h is the length of the dipole 

element, A is the area of the current loop,and N is the number of 

turns in the loop. 

Taking the ratio of the open circuit output voltage induced in 

the receive antenna of each pair of like antennas (i.e., dipole-dipole, 

loop-loop), we get the comparison factor 

3 

Voc (dipoles) ~~h~ 

F = - 

d,~ Voc(lOOps) -2 N I~A 2 wpa ' 

Substituting into (11) the representative values of h = 0.5 m, A = 0.2 m 

2 

based on a 0.5 m loop diameter, N = 10 turns, p = po = 4rr x 10-~h/m, 

-2 conductivity a = 10 mho/m, and frequency f = 300 kHz, we find that 

F reduces to 

d,R 

F d,. =(?) 

Therefore, it becomes apparent that based on this measure of performance 

for antennas in a conducting medium, the advantage enjoyed by the dipole 

over the loop in free space does not apply in a conducting medium. A 

Arthur D Little Inc. 
similar conclusion was reached by investigators (I5' 16)at the University 

of Innsbruck in Austria during the course of a VLF through-the-earth 

communications project funded by the U. S. Air Force. Furthermore, 

when the difficulty of driving the antennas to obtain the required 

currents I and I& is also factored into this comparison equation, d 

the loop antenna is clearly seen to be the most favorable ante-. 

For example, the resonant impedance of a tuned loop having a 

bandwidth of 12 kHz at 300 kHz is about 10 ohms, whereas the practical 

grounded dipole source is faced with the spreading resistance of its 

two ground connections to the medium. These two connections can 

take on values greater than 200 ohms, based on a simple 0.5 m ground 

-2 rod termination at each end of the dipole in a medium having a = 10 mho/m. 

If both antennas are driven by the same voltage, the resultant performance 

factor F is given by: d,R 

Substituting the above parameter values, and taking into account the 

increase in the effective length of the dipole antenna caused by the 

insulated ground rod terminations, we find that F = 0.1. Since the 

d,R 

medium conductivity a also appears in the numerator of F when Zd is 

d,R 

the spreading resistance of ground rods, this performance comparison 

should remain insensitive to the conductivity of the medium for the 

conductivities and frequency band of interest. The lower resonant 

impedance of the loop will also allow better receiver utilization of 

the induced open circuit voltage. Thus, the loop exhibits a clear 

performance advantage over a typical grounded-dipole antenna. 

The advantage of the loop becomes greater when compared to a 

whip-type dipole antenna,(6) which would be the practical choice over 

the grounded dipole for a roving miner's portable radio. This whip 

would be a completely insulated antenna not grounded to the medium at 

each end. The impedance of this antenna worn by a miner standing in 

a tunnel will be a high capacitive reactance instead of the modest 

resistance of the grounded dipole. The spatial coupling to the 

surrounding medium would also be reduced. 

I 

Arthur D Little, Inc 
Examining the impact of the input impedance change, we find that 

the capacitance represented by the short whip worn on a person is very 

low (approximately 10 pf), and subject to large variations (17918) due 

to "stray" capacitances such as those introduced by the whip's variable 

proximity to the body and mine environment and by the large tuning coils 

required to resonate such a small capacitance at 300 kHz. This 

situation typically requires that a lumped capacitance much larger than 

the antenna capacitance be placed in parallel with the antenna, to 

allow the use of a practically sized tuning inductor at 300 kHz, and 

to prevent changes in the resonant frequency caused by variations in 

the effective antenna capacitance. This lower reactance shunt path 

will divert most of the tuned circuit current into the large shunt 

capacitor instead of the high reactance antenna. Antenna current 

reductions of 100 to 1 are not uncommon, thereby putting the short 

whip-type dipole antenna at a considerable practical disadvantage for 

the present roving miner application. The following sections of this 

chapter discuss the feasibility and potential utility of specific 

antenna types . 

C. PORTABLE MANPACK ANTENNAS 

1. Conventional Whip Antennas 

The objective for the portable mine wireless application is to 

obtain an antenna that is not only effective but small, rugged, and 

convenient for a miner to wear. Electric dipoles or whip antennas fail 

tomeet this objective in all respects at the desired operating frequencies in the mine environment. Therefore, only a brief review of 

some of the major shortcomings is given below. 

The size, convenience, and ruggedness problem can be appreciated 

by comparison with HF ba.nd manpack hardware used for tactical radio 

comunications by the military, a user which generally attempts to 

optimize the same characteristics of concern to a mine wireless 

application. A case in point is the PRC-74 transceiver which requires 

a 6-foot, inductively loaded whip to achieve reasonable performance on 

the surface over most of the HF band. (9) However, the performance of 

even such whips, with inductive loading optimally placed about 0.4 of 

Arthur D Little Inc. 
the way up the whip, falls off rapidly below about 6 MHz when the whip 

becomes less than 0.05 A. (lo) A semi-f lexible whip structure of this 

type and length is totally incompat ible with underground roving miner 

mine wireless applications. The military also considers it highly 

inconvenient, particularly so in jungles, but tolerates it because of 

the practical necessities of tactical military communications on the 

surface. The magnitude of the problem of course becomes much more 

severe at frequencies in the MF and lower bands of interest for mine 

wireless radio, particularly when a miner is likely to consider even 

a 25 cm whip unreasonable. 

Whip effectiveness with respect to signal strength produced in a 

conducting medium and the ease of generating such a signal from a 

practical transmitter is also poor. The previous section of this 

chapter showed that ideal loop antennas are more effective than ideal 

grounded-dipole antennas in a conducting medium at the operating 

frequencies of interest. The problems and performance associated with 

whips rapidly worsen for portable manpack applications, making loop 

antennas the clear choice for portable mine wireless applications. 

As mentioned briefly in Section III.B.8 an ideal electrically 

small whip of about 1 meter length over a perfect ground plane is 

characterized by a very small capacitance of approximately 10 pf, (17) 

which results in a very high capacitive reactance at frequencies in the 

100 kHz to.1 MHz band. The tuning difficulties imposed by this whip 

antenna, even in such a simple environment, become apparent by looking 

at the experience at HF with 6-foot whips; namely, they are nearly 

impossible to resonate with a practical series lumped inductor below 

6 MHz because of the whip's high capacitive reactance. (I7) For 

example, a 10 mh inductor is required to resonate a 10 pf capacitor at 

300 kHz. The problems worsen when such a whip must be worn on the body 

of a person in a manpack application. The whip's geometry, and its 

high capacitive reactance below low band VHF, make the whip's input 

impedance strongly affected by the whip's closeness to and position on 

a person's body, and by the person's immediate local environment, his 

position within it, and his "connection" to it via his feet or other 

parts of the body. (17,18,1.9) 

Arthur D Little, Inc 
The practical problems associated with tuning and driving the high 

and widely variable input impedance of such an antenna configuration 

become intolerable. Thus, it becomes desirable, if not necessary, in 

practice to: either set the unit down away from the body and surrounding 

objects; or incorporate circuit elements that minimize the antenna 

impedance variations seen by the transmitter and receiver and at the 

same time simplify the tuning coil design. Only the latter option is 

operationally allowable for the MF mine wireless application. A 

typical means of accomplishing this latter option is by adding a large, 

lumped shunt capacitor in parallel with the whip. This of course 

shunts a large portion of the transmitter output current away from the 

whip, leading to the correspondingly large reduction in overall system 

effectiveness discussed in Section 111. B. 8. 

In summary, problems associated with whip antennas for use in 

portable manpack applications are not new, nor are they unique to the 

mine wireless application. They have been and continue to be a source 

of considerable concern and difficulty at both low band VHF and HF, 

with the problems becoming particularly acute at the low end of the HF 

band. Thus, it is n o t . s u r p r i s i n g that conventional whip antennas are 

incompatible with the MF band mine wireless radio application. 

2. Active Whip Antennas 

A moderate amount of attention has been given since the mid-1960's 

to investigations of the potential advantages of so-called active 

antennas which incorporate a transistor into the whip antenna structure 

itself. Investigations have included the frequency range from VLF to 

UHF, and both receiving and transmitting applications. (20-29) 

Receiving applications got most of the attention until the early 1970's 

when effort devoted to transmit applications was increased. (20,21,22,23,24) 

The intent of this section is not to present a detailed explanation and 

assessment of this work, but to briefly describe and comment on an 

active transmitting configuration that has been reported (20,21) to 

offer certain performance advantages over that of a conventional seriestuned whip antenna in the MF band. Our conclusion is that this active 

antenna configuration does not overcome the considerable disadvantages 

Arthur D Little, Inc. 
associated with whip-type antennas for portable mine wireless applications, and moreover it Is significantly more wasteful of power than 

the conventional tuned whip. 

Figure 3-7 depicts the fed-emitter base loop (FEBL) transmitting 

antenna configuration which is considered by active antenna investigators (20'21) to be favorable for transmitting applications. Z is the 

S 

collector bias resistor connected to the DC power supply. The optimum 

transistor position for a transmit application is the one that maximizes 

the current moment Ihof the antenna. This occurs for the FEBL configuration of Figure 3-7 when the transistor is at ground plane level (20) 

as shown in Figure 3-8. This transistor position maximizes the 

collector and emitter currents, and also eliminates radiation field 

cancellation caused by rf currents flowing in the negative direction in 

the bias resistor lead. By using a 3000 ohm collector bias resistor of 

value much less than the antenna capacitive reactance (for C " 7 pf), A 

the investigators obtain a relatively frequency independent resistive 

input impedance of about 30 ohms at the emitter-base terminals of a 

1 meter long transistorized antenna. Thus, "so-called" resonant 

operation at a real input impedance is obtained with an electrically 

short,capacitively reactive antenna without using a tuning inductor. 

Furthermore, the impedance seen by the transmitter is buffered from 

variations in antenna reactance caused by the environment and frequency 

over a wide range in the MF and HF bands. 

However, the bias resistor in parallel with the whip antenna's 

high capacitive reactance provides a low impedance shunt path for the 

collector current as shown in the equivalent circuit of Figure 3-9a, 

thereby diverting most of the available current away from the antenna as 

in the case of capacitance shunting of conventional whips discussed 

previously. In addition, this shunting resistor greatly increases the 

power dissipation of the total antenna circuit over that required to 

produce the same antenna current in a series-tuned antenna equivalent 

circuit of Figure 3-9b. 

In reference (20), the above described active antenna configuration 

is compared at a frequency of 1.24 MHz with a series-tuned antenna 

It 

Arthur D Little, Inc 
Source: Reference 20. 

FIGURE 3-7 ACTIVE TRANSMIT ANTENNA 

FEBL CONFIGURATION 

Source: Reference 20. 

FIGURE 3-8 FEBL CONFIGURATION FOR 

MAXIMIZING ANTENNA 

CURRENT MOMENT 

(a) 

Equivalent Circuit For FEBL Transmit Configuration Of Figure 3-8 

w - 

(b) 

Equivalent Circuit For Series Tuned Conventional Whip 

Source: Arthur D. Little, Inc. 

FIGURE 3-9 EQUIVALENT CIRCUITS 

Arthur D Little Inc. 
circuit having a Q of about 120 (10 kHz bandwidth). By neglecting the 

power dissipated in the bias resistor, the active antenna is shown to 

exhibit a "so-called" power gain disadvantage of about 9 dB compared to 

the tuned antenna circuit. If account is also taken of the power 

dissipated in the bias resistor, this disadvantage increases considerably 

to about 27 dB; namely, the active whip antenna dissipates at least 

27 dB more power than an equivalent series-tuned whip antenna. 

Finally, the antenna current values obtained from the active 

antenna configuration examined in reference (20) are characteristic of 

the levels obtainable from a 12 volt battery suitable for a miner's 

portable radio. These values of 0.4 ma at 1.2 MHz and 0.1 ma at 300 kHz 

are clearly inadequate, except perhaps for extremely short range, nearfield applications in free space. Thus, although active antennas may 

offer certain performance advantages over conventional whips in some 

applications, they offer no practical benefits as transmit antennas for 

MF portable mine wireless radios. 

3. Conventional Loop Antennas 

Conventional air-core loop antennas represent one of the most suitable 

and effective choices forthe MF mine wireless roving miner application. 

Thus, it is not surprising that the South Africans (30931) and the 

British (32,33,34 &5) have adopted the use of such antennas for rescue 

type, portable MF band communications in mining and fire fighting applications respectively. Similar developments are also occurring within 

the U. S. coal mining community. (36,37,38,39,40) However, the stateof-the-art is such that, even with such loops, a completely wireless 

range goal of 1350 feet in coal mines is likely to be attainable only 

in the most favorable mine environments (41y42943) (see Chapter 11). On 

the other hand, ranges on the order of 1 - 3 km have been reported (30, 

31935936y40944945)when such loop antennas are employed in the vicinity 

of mine electrical conductors (see Chapter IV for representative calculations) . The conventional loop antenna has a large number of positive 

attributes for roving miner applications. At the low frequencies of 

interest it is an inductively reactive device whose magnetic fields 

I 

Arthur D Little, Inc 
and input impedance remain, for all practical purposes, unaffected by 

its location on and closeness to the body of a person, and also by the 

local mine environment. The stray capacitance problems associated with 

whip antennas are avoided. The moderate inductive reactance of a typical 

loop configuration, having on the order of 10 turns, can be easily 

tuned by readily available practical capacitors in the MF band. The 

loop can be readily formed into a lightweight, rigid or flexible, 

bandolier shape. This allows the area of the loop, and therefore the 

transmit magnetic moment, to be maximized within a form factor that can 

be tailored to protect the loop from damage while providing the least 

inconvenience to the miner wearing it. Some examples of current designs 

are presented below. 

Rigid units like the tubular elliptically shaped bandoliers 

depicted in Figure 3-10aare used with the South African radio (30) (SAR) 

at 335 kHz and the Collins Radio mine wireless radio prototypes (37) at 

520 kHz. They provide the advantage of a fixed area, and therefore 

fixed transmit moment and loop inductance,butthe disadvantage of somewhat greater awkwardness than flexible bandolier conf igurations . Representative flexible units include: the strap-type depicted in Figure 

3-lob used by Plessey in its Inductorfone TGR.1 unit (35) at 140 kHz and 

by Lee Engineering Division of Consolidation Coal Co. in prototype 

units (36) at 425 kHz, and garment-type depicted in Figure 3-10c where 

the turns (typically stranded wire) are sewn in the desired shape into 

the person's work uniform or into a specially designed jacket or pullover worn by the radio-equipped person. For example, the Plessey 

#PRD2200 unit used by firefighters in large buildings at 2.5 to 3.5 MHz 

is incorporated into a special jerkin, (33) whereas one of the Lee 

Engineering prototype units (36) at 425 kHz has the loop sewn into a 

pair of miner's coveralls approximately as shown in Figure 3-10c. The 

Rus s ians (45) have also made use of loop antennas, in a hand-carried 

configuration, for mine radio communications at 78 kHz. The loop 

orientation which results from wearing a bandolier type of antenna 

(i.e., in a nearly vertical plane) also closely approximates the ideal 

orientation for coupling to the quasi-TEM mode of propagation in the 

Arthur D Little Inc 
a. Rigid Oval Tubular Type b. Flexible 

Source: References 30, 35, 36, 37 and Arthur D. Little, Inc. 

FIGURE 3-1 0 SKETCHES O
coal seam waveguide (see Figure 2-6). The following paragraphs are 

devoted to examining some design guidelines and performance limits of 

such antennas for a portable mine wireless radio application. 

The first design objective is to generate the largest practical 

antenna magnetic moment within the constraints imposed by intrinsic 

safety, reasonableness of size and form factor for wear by a miner, and 

practical tuning and matching elements. The second and equally important design objective is to produce this magnetic moment by the most 

efficient use of the energy stored in the battery of the portable unit. 

The first objective affects the maximum attainable range, while the 

second affects the maximum allowable operating time between battery 

recharges, or conversely, the minimum size battery required to obtain 

the minimum desired operating time (such as a normal work shift). 

Furthermore, the signal attenuation and electromagnetic noise encountered in typical coal mines is so severe that attempts to achieve smallto-moderate increases in transmit magnetic moment, at the expense of 

overall transmitter efficiency or power dissipation, are generally not 

warranted. In a roving miner application, it is better to have a 

smaller, lighter-weight, longer-lasting unit that has a somewhat shorter 

range than a unit that has 100 - 200 feet greater range under some 

highly favorable mine conditions, but is too heavy and bulky for the 

miner to wear, or has too short an operating time between recharges. 

The transmit moment is maximized by maximizing the product M = NfA 

of the loop, where N is the number of turns, 1 is the current flowing 

in the turns, and A is the area enclosed by the turns of wire. The 

maximum reasonable loop area for a roving miner application is about 

2 0.25 m . Both the SAR and the Collins Radio prototype MF units (30,371 

use a rigid tubular elliptically shaped bandolier (SAR - 0.66 m x 

0.42 m, Collins Radio - 0.61 m x .47 m, 2 ft. x 1 ft.) having an area 

2 about 0.22 m . The Lee Engineering units(36) use a strap-type bandolier 

of dimensions 0.56 m x .33 m, and a garment-sewn circular loop having 

the same perimeter as the elliptical strap unit and an area of about 

2 0.15 m . 

Arthur D Little Inc 
A practical number of turns appears to be in the 5 to 15 turn range 

(SAR and Collins - 7 turns, Lee Engineering - 12 turns). The inductance 

of an aircore loop having a diameter d and a single layer winding of 

width w is given by the equation 

L M~'~N~ microhenries (14) 

from Watt,@) where A is the loop area in square meters, N is the 

number of turns, and k is the inductance form factor which depends on 

the ratio of d/w. k is a slowly changing function for the d/w range of 

present interest, varying from a value of about 1.8 for d/w = 5 to a value 

2 of about 3.6 for d/w = 50. Thus, a loop having an area A = 0.25 m , 

10 turns, and an effective d/w ratio of about 25 (k = 3) results in an 

inductance of about 150 ph, which can be easily series-tuned with practical capacitor values on the order of 1000 pf in the 300 - 500 kHz 

frequency band. Reducing the turns to 7 gives about 75 ph, which is 

close to the measured values of 60 ph and 82 ph for the SAR and Collins 

loop antennas, respectively. Thus, the capacitor values required for 

tuning increase accordingly to between 1000 pf and 5000 pf, depending on 

the inductance value and operating frequency. 

A bandwidth of 12 kHz in the 300 to 500 kHz band requires a Q of 

2 25 to 42 respectively. For a series-tuned 8-turn, 0.25 m , loop of 

nominal 100 ph inductance, this means a real impedance level of 7.5 ohms 

at the resonant operating frequency. Thus, an rms loop current on the 

order of 1 ampere is obtainable from a transmitter operating from a 12 

volt battery supply. A double totem pole, class D, transistor bridge 

output stage is one example of a highly efficient way to utilize the 

full battery voltage on both positive and negative half cycles. This 

allows the generation of a practical rms transmit moment, M = NIA, of 

2 about 2 amp-m , a value nearly equal to that of the Collins Radio 

prototype unit, and slightly higher than the moment of the SAR unit. 

The intrinsic safety curve shown in Figure 3-11 (46,47,48) is 

based on the findings of the Mining Enforcement and Safety Administration 

(MESA) for DC inductive circuits. It reveals that for an inductance 

of 100 ph the rms current of 1 ampere is already close to the band 

lli 

Arthur D Little. Inc 
100 ma 1 a 

Current 

Source: Ref. 46. 

FlGURE 3-11 MINIMUM IGNITING CURRENT AS A FUNCTION OF ' 

AN INDUCTIVE CIRCUIT FOR AN 8.3% METHANEIAIR 

MIXTURE 

Arthur D Little Inc 
defining the zone of maximum safe current level. In AC applications, 

this level must represent the peak current (48) level, so the 1 amp rms 

level compared to the allowed 2 amps peak includes a modest safety 

factor. It can be seen from the plot that significant gains in transmit 

moment cannot be obtained from increases in current at this inductance. 

Nor can they be obtained by increasing the number of turns, since the 

inductance, and also the resistance for a fixed bandwidth requirement, 

2 increases more rapidly (as N ) than the moment (as N), thereby reducing 

the maximum attainable and intrinsically safe current. 

In the other direction, namely reducing turns, the current at 

resonance will increase as the inductance and therefore the resistance 

decrease. To avoid problems with stray circuit and cabling inductance, 

the loop inductance should be kept above about 10 ph. This allows a 

3 to 1 reduction in turns, a 9 to 1 reduction in L and R, a corresponding 9 to 1 increase in I, and a net increase of 3 to 1 in magnetic 

moment M. However, even this modest increase in M is unrealizable in 

practice. The intrinsic safety curve becomes increasingly steep below 

1 mh and particularly below 100 ph, (46 948) thereby allowing at most 

perhaps a 1.5 to 1 increase in current, which results in a net decrease 

in M instead of an increase. The practically achievable moment for a 

7-turn loop having an inductance of about 75 ph, like the SAR and 

Collins loops, will be about the same as that for the 8-turn, 100 uh 

loop, and the 1 amp rms in the 7.5 ohm load will dissipate 7.5 watts. 

The rf voltage across the tuning capacitor (and the loop inductor) 

at an operating frequency of 500 kHz will be as high as 318 volts rms 

(450 V-peak) for a 1000 pf capacitor carrying a current of 1 amp rms. 

These voltages fall well within the safe area of the intrinsic safety 

plot for capacitors (46) shown in Figure 3-12. 

It should also be realized that existing intrinsic safety curves 

are based on DC circuit experience. The subject becomes much more 

complex (48) for AC applications, so the DC-based curves serve only as 

approximate design guidelines for the MF band radio application. In 

the end, verification of the MF mine wireless portable radio's intrinsic safety will be determined only after the radio, including the 

Arthur D Little Int 
FIGURE 3-12 MINIMUM IGNITION VOLTAGE AS A FUNCTION OF 

CAPACITANCE FOR IGNITION OF AN 8.3% METHANEIAIR 

MIXTURE 

Arthur D Little Inc 
antenna, has successfully completed methane ignition tests conducted by 

MESA in its special test facility. 

The above discussion shows that practical values of magnetic 

moment M for a portable MF band unit are not likely to exceed the values 

2 of about 2.5 amp-m peak available with present prototype equipment. 

This represents only a 12 dB improvement in signal levels over those 

obtained in the performance estimates of Chapter I1 for a moment of 

2 0.7 amp-m . In highly favorable mine attenuation and noise environments, 

this may result in a range increase of about 300 ft. over that predicted 

in Table 2-1 of Chapter 11. In fair-to-poor mine environments, range 

increases of less than 50 ft. will occur and the completely-wireless 

communication range will remain hopelessly small in these situations 

as shown in Chapter I1 and reference (41). 

The major objective for MF mine wireless radios should therefore be 

to obtain the most reasonable value of magnetic moment consistent with 

a transmitter/antenna circuit design that provides the most efficient 

use of the battery's energy supply. This will lead to a portable 

system with the smallest battery size, and therefore smallest portable 

unit size and weight, for satisfying the principal operational requirements, of which the wearability and operating lifetime between charges 

are as important, if not more so, than the magnitude of the transmit 

magnetic moment. This suggests the use of MF transmitters designed to 

operate in a large-signal mode (such as the one mentioned above) to 

obtain efficiency benefits similar to those of switching-type transmitters at lower frequencies, and optimized for driving a portable 

unit antenna having a relatively low tuned impedance at the operating 

frequency. 

In summary, air-core , bandolier-type loops provide a feasible , 

practical solution to the MI? wireless radio antenna problem, but at a 

price of minor-to-moderate inconvenience to the miner, depending on 

whether the antenna is sewn into a garment or fashioned into a flexible 

or rigid bandolier. The most suitable and practical configuration for 

a particular mine application is best determined by in-mine trials 

with prototype designs. The convenience of wear offered by the garment 

Arthur D Little, Inc 
and flexible bandolier configurations will be somewhat offset by the 

smaller and slightly variable transmit moments and inductances. The 

inductance variation could possibly, but not necessarily, introduce 

the inconvenience of minor retuning of the antenna on different sized 

individuals. This can be easily checked by experiment. Finally, the 

transmit moments of presently available prototype units are representative of what is attainable within the operational constraints of the 

coal mine MF wireless application; and overall portable system improvements will most likely take the form of increased efficiency of transmitter utilization of battery energy, and consequently reduced overall 

size and weight of the portable units. 

4. Ferrite-Loaded Loops 

In many applications, loops loaded with magnetic core materials 

offer equal or improved performance in a considerably reduced crosssectional area and volume over that of an air-core loop. (6) Perhaps 

the most familiar example of such an antenna in the MF band is the 

smal1,ferrite loop-stick receiving antenna used in small transistor 

pocket radios. A typical configuration for many ferromagnetic core 

loop antenna applications is shown in Figure 3-13. (6) I£ such an 

antenna could provide the required magnetic moment within a cylindrical 

volume of 10 in. length by 2 in. diameter, it would become quite 

attractive for a portable mine wireless radio, because it could then be 

mounted horizontally to the radio unit itself or to the miner's belt on 

which the cap lamp battery and self-rescuer are attached. The brief 

feasibility calculation described in the following paragraphs shows that 

the practically attainable transmit moment for such a ferrite-core loop 

compares favorably with that obtainable with an air-core bandolier loop. 

Thus, experimental effort should be devoted to verifying its feasibility, 

and if confirmed, to establishing the optimum antenna design for a 

portable mine wireless radio. 

When a loop loaded with a high-permeability ferrite core or rod, 

as shown in Figure 3-13, is placed in an ambient magnetic field, the 

magnetic flux passing through the cross-sectional area enclosed by the 

loop is greatly increased. (6 911949 350) The high-permeability rod, in 

Arthur D Little Inc 
Source: Ref. 6 

FIGURE 3-13 TYPICAL CONFIGURATION FOR LOOPS LOADED 

WITH FERROMAGNETIC CORES 

Ferrite Coil 

Rod Core 

Source: Ref. 1 1. 

Source: Ref. 1 1. 

FIGURE 3-14 SKETCH OF FLUX CONCENTRATION 

CAUSED BY HIGH PERMEABI LlTY ROD 

FIGURE 3-15 APPARENT PERMEABILIT

FUNCTION OF ROD P/d A

PERMEABILITY, p 
effect, gathers and concentrates the magnetic flux lines in the vicinty 

.of the rod, and causes them to pass through the rod which threads the 

loop, as depicted in Figure 3-14. This greatly increases the "captured" 

flux, the "effective area," and therefore the induced open circuit output voltage of the loop acting as a receive antenna. The presence of 

the rod can also be viewed as increasing the effective permeability of 

the medium surrounding the antenna, thereby creating the increased flux 

linkage and output voltage of the loop. By reciprocity, it can be shown 

that the magnitude of the fields generated by such a loop, when acting 

as a transmit antenna, will be similarly a rod of magnetic 

material. The transmit improvement will 

reception within the linear operating region of the material. In practice, significant improvements in transmit magnetic moment can be realized with practical materials until the materials are driven into 

saturation. . 

The presence of the magnetic rod core causes the flux linked and 

produced by the loop, the loop inductance, and the loop Q, to be increased 

by different amounts. The amount of increase depends on the shape and 

size of the rod and on the relative permeability, v, of the rod material, 

also referred to as the intrinsic permeability of the rod material when 

measured in a toroid (utoroid ) ( l1 A considerable amount of analytical and experimental work has been done (6 9 11 ,50,51,52,53) to quantitatively 

describe how the above quantities depend on the intrinsic permeability 

of the rod material, on the shape or R/d ratio of the rod, and on the 

length of the rod compared to the length of the loop or coil, namely 

the R/W ratio. 

Referring to Figure 3-14, the best overall practical performance 

of ferrite rod antennas generally occurs when &/d 5 20, t/w 8, and 

R/D s 4. Furthermore, as illustrated in Figure 3-15, when the intrinsic 

permeability p is greater than about 500 to 600, the effective permeability of the rod urod becomes essentially independent of the intrinsic 

This is particularly so for the R/d range of principal interest for 

the MF mine wireless radio, namely L/d 6 20. Figure 3-15 shows the 

apparent or effective permeability of the rod, (increase in 

Arthur D Little Inc 
received flux or transmitted field strength for the loop) as a function 

of length-to-diameter ratio, &Id, of cylindrical rods having various 

intrinsic or toroidal permeabilities, v. The increase in the inductance 

of the coil or loop caused by the rod can be expressed as an increase 

in the coil permeability, v coil The increase in vcoil is generally 

much less than the increase in v-_,, except when the &Id ratio is not 

(6,111 LUU 

much larger than unity, and "oil generally increases with the 

length of the coil on the rod. w.Q of the coil is also increased by 

. the presence of the rod, so the rod material pQ product in the frequency 

band of interest is generally an important design parameter. ' A 

plot of typical pQ product behavior as a function of frequency below 

10 MHz can be found in reference (51). 

To assess the feasibility of using a ferrite rod loop antenna for 

the portable mine wireless MF radio, a rod material form factor and 

size is chosen that offers favorable performance, is compact enough to 

be conveniently worn by a miner, and is readily available from suppliers. 

A 10 in. long, 1 in. diameter (&/d - 10) rod of manganese-zinc (MnZn) 

ferrite material typically used for antenna rods in the LF and MF bands 

from about 100 kHz to 2 MHz is assumed. (54 955) This material typically 

baa a saturation flux density B of about 0.3 tesla and an intrinsic sat 

initial permeability p of 500 to 1000. 

The saturation flux density defines the maximum moment that a 

given volume of the material can produce. This maximum M is given sat 

by 

when the flux density is uniformly distributed over the volume. Bsat 

is the saturation flux density in teslas, V is the volume of the rod 

3 -7 in m , and p = 47~ x 10 h/m. Substituting for B sat' 'rod' and p we 

0 0) 3 

find that the maximum attainable moment for the rod is M = 31 amp-mL sat 

peak, a value greater than 10 times the maximum intrinsically safe 

moment obtainable with a practical bandolier air-core antenna. However, 

as shown below, it is not possible to generate this maximum moment in 

an intrinsically safe manner or with a practical 12 volt transmitter. 

C* 

Arthur D Little Inc. 
solenoidal coil with this form factorsthe simple air-core inductance 

formula applies (56) 

2 2 

= a microhenries, Lo 9a + IOW 

which then must be multiplied by pcoil (6*11) to obtain the inductance 

when it is ferrite loaded. 

N is the number of,turns, a is the coil radius in inches, and w is the 

coil length in inches. The value of p coil = 10 is used, based on the 

finding that for rods having an L/d ratio of 12, p (52,6) 

coil ' prod11o* 

Thus, substituting for u coil' a, w, and N in (17) and (18) gives a ferriteloaded loop inductance of L = 125 ph, a value close to that obtained 

with an 8-turn air-core bandolier antenna. 

The bandw'idth of 12 kffi at an operating frequency of 500 kHz 

rcqu2res a Q of 42. To achieve this Q at 500 kHz, a series-tuned real 

Arthur D Little lnc 
impedance level (R = w~/~)of 6.7 ohms is requir d. This results in a 

slightly lower peak current of about 1.2 amper a s than the 1.4 amperes 

assumed above for estimating the number of tur (N = 35) required to 

produce the desired magnetic moment of 2.5 amp 

Therefore, we find that this ferrite-load d loop configuration is 

nearly equivalent to the 8-turn bandolier loop configuration of the 

previous section in all respects, including it location inductance and 

current-wise with respect to the safe region oundary on the intrinsic 

safety plot of Figure 3-11. As in the case of he bandolier loop, a 

modest decrease in the number of turns allows moment to be optimized 

with respect to the intrinsically safe current inductance levels. In 4 

this case about 28 turns should provide the de ired peak moment of 2.5 

amp-m2 at the inductance and peak current leve s of about 80 ph and 

1.8 amperes respectively. Similar results can also be achieved by 

distributing the 35 turns over a greater lengt 

The above example also illustrates that t e maximum achievable P 

moment M defined by the saturation flux d sity B and volume of sat' T sat 

the ferrite rod, would require current levels in excess of those which 

are intrinsically-safe and greater than the 12 volt power supply could 

supply to the real impedance level imposed by he bandwidth requirement. 

As in the case of the air-core bandolier loop,) the maximum intrinsically2 safe moment is constrained to be about 2.5 amplm . However, the prospect 

2 of achieving the M = 2.5 amp-m performance wi h a much smaller antenna f 

than the bandolier loop makes the ferrite-load d loop very attractive I= 

for the roving miner application. In fact, example suggests that 

this moment could be achieved with even a volume core than the 

one used in the example. The favorable performance of ferrite loaded 

loops as receiving antennas (6911951) make their use doubly attractive. 

Therefore, effort should be directed toward verifying this predicted 

performance. This is best accomplished e~perimentally, as will the future optimization of any practical design, should the feasibility be 

confirmed. 

5. Special Multiturn Loop (MTL) Antennas 

The Ohio State University and others have been studying and 

developing a class of electrically small antennas called multiturn 

loop (MTL) antennas. (7,57,58,59,60,61,62) The objective has been to 

Arthur D Little Inc. 
pr~vide improved performance in terms of antenna system radiation 

efficiency, or comparable performance in a smaller and more convenient 

volume, than that obtainable with whips tuned by generally high-loss 

coils in the HF band and in the low and high VHF bands. Althoilgh these 

MTL antennas may offer size and/or performance advantages for some 

applications in the HF and VHF bands, they are by and large inappropriate 

for the portable mine wireless MF band application. 

MTL antennas are low-loss, high-Q, single-layer coil structures 

having ~/d ratios near unity, and consisting of from 3 to 10 turns of 

tubular or strip conductor, spaced about 2 to 4 conductor diameters 

apart. The axis of the MTL coil is positioned parallel to, and at a 

short height above,a ground plane. Tuning and matching is accomplished 

with two low-loss capacitors. As depicted by the antenna schematic in 

Figure 3-16a, the coil's self-capacitance between turns, and between 

the turns and the ground plane, is supplemented by the parallel tuning 

capacitor C to produce a parallel resonance at a frequency, A fr, above 

the desired operating frequency, f . Below fr, the parallel combination 

0 

of C and the coil structure exhibits an input impedance having real and A 

inductively reactive parts as shown in Figure 3-16b. This impedance is 

in series with the capacitive reactance of the matching capacitor, CB. 

To obtain the desired value of real input impedance, the value of CA 

is adjusted, which shifts f so as to produce the desired real part of 

r 

the parallel coil-capacitor circuit impedance at the operating frequency, 

f . Then, the matching capacitor C is adjusted to cancel the inductive 

0 B 

reactance of the antenna structure at f . The result is a high-Q, 0 

narrow bandwidth, tuned antenna structure capable of moderate-to-high 

radiation efficiencies. Best results are generally obtained when the 

largest structure dimension is greater than at least XI50 (preferably 

greater than X/20) and the length of the conductor forming the turns 

is between X/8 to X/4. 

Two examples of portable MTL applications are the prototype packset 

MTL antenna developed for the PRC-77 low-band VHF (30-76 MHz) military 

transceiver (57 958) and a prototype shoulder-mounted MTL antenna for 

high-band VHF police personal radios. (61) The size of each MTL would 

Arthur D Little Inc 
a. Schematic Diagram of MTL Antenna 

b. Impedance Characteristics of MTL Antenna 

Source: Reference 7. 

FIGURE 3-16 MTL ANTENNA DIAGRAM AND IMPEDANCE CHARACTERISTICS 

Y 

Arthur D Little Inc. 
be compatible with a portable mine wireless application. The low-band 

VHF packset MTL is a 4-turn square coil of dimensions 2-1/2 x 2-1/2 x 

4-1/2 inches long, mounted to the base of the PRC-77 radio, and enclosed 

in a fiberglas protective cover which extends 5-1/2 inches below the 

packset, which is carried as a backpack by a soldier. This MTL was 

found to compare favorably to a 3 ft. whip over the 30-76 MHz band, but 

to also experience tuning and pattern problems as a result of the 

antenna's high Q and its closeness to the body. The high band VHF 

shoulder-mounted MTL is 2.7 x 2.7 x 0.7 inches high, mounted above a 

4.5 x 6.5 inches shoulder-pad type of ground plane. This MTL was found 

to compare favorably with a 6 inch helical whip over the 150-170 MHz 

band, but to also experience problems similar to those for the low-band 

VHF packset MTL. Both required tuning after being mounted on the 

person, and the performance of both depended on body positions and 

orientations. At MF band wavelengths, which are between 2 and 3 orders 

of magnitude greater than those at VHF, the problems associated with 

portable-sized MTL units will become significantly more severe. 

In the VLF to MF band of present interest, the MTL dimension requirements with respect to wavelength are clearly incompatible with the 

sizes desired for portable mine wireless units, and even for semipermanent fixed installations in coal mines. The ground plane requirement is similarly incompatible. Furthermore, as cited in Section III.B, 

high antenna efficiency requires high antenna system Q, which is also 

incompatible with the bandwidth required for mine wireless FM voice 

communications in the VLF to MF band. Thus, the principal attribute 

of ideal MTL antennas is neither desired nor attainable for the mine 

wireless application. Therefore, although MTL antennas may offer 

performance advantages over other antennas for portable radio application~'~~'~~ '61) in the VHF band, and for shipboard applications (7) 

in the HF band, MTL antennas are inappropriate for MF band mine wireless 

applications. 

Arthur D Little. Inc 
D. FIXED STATION ANTENNAS 

This section briefly considers some alternative antenna types and 

radio wave coupling methods that are suited to mine wireless radio fixed 

installations such as base stations or repeaters. 

1. Vertical Plane Loops 

Air-core planar loops, series tuned, provide a practical choice 

for the mine wireless radio portable units. They also provide the most 

convenient way of generating the largest magnetic moment per unit weight 

of antenna for a fixed radio installation. This is especially so when 

space is available to span a large area with the loop as is usually the 

case in coal-mine tunnels. The planar loop combines the desirable 

attributes of high magnetic moment, well-defined and stable impedance 

characteristics, and ease of installation and maintenance. 

A practical loop configuration for a fixed installation is a 

rectangular air-core loop, series tuned, and formed in a vertical plane 

so that it can be "hung" or mounted on the wall or rib of the mine tunnel 

as depicted in Figure 3-17. The vertical orientation of the planar 

antenna is the most favorable for exciting the desired zero-order, quasiTEM mode having horizontal magnetic field H and vertical electric field @ 

E in coal seams (Section II.D, Figure 2-6). Maximum signal strengths z 

will occur at positions in the plane of the loop (4 = oO), and minimum 

signal strengths at locations in the vertical plane perpendicular to, 

and passing through the center of, the loop ($ = 90°), according to the 

cos $ angular dependence of a magnetic dipole (loop) source. 

The angular dependence of the mode excitation presents a nonuniform 

area coverage problem that is somewhat more serious than the one 

experienced by a miner wearing a bandolier-type loop antenna. For 

example, the miner can always rotate his body position to maximize his 

received signal-to-noise ratio. However, the same result can be obtained 

with fixed station installations by installing a second vertical planar 

loop oriented perpendicular to the first loop, and driven with a current 

equal to, and 90'. out of phase with, the current in the first loop as 

shown in Figure 3-18. This is a turnstyle type of antenna installation (11) 

Arthur D Little Inc. 
lo cos (at - 77/21 I lo cos at 

Source: Arthur D. Little, Inc. 

FIGURE 3-17 WALL-MOUNTED FIXED STATION LOOP IN 

COAL MlNE TUNNEL 

a. Ideal Turnsty le-Ty pe 

Antenna Configuration 

L Resultant Rotating 

Source Strength 

b. Top View Turnstyle 

Configuration 

I 

h0 

c. Equivalent Mechanically 

Rotated Antenna 

lo cos (at - 

Source: Arthur D. Little, Inc. 

a r,l 

\- 

\lo COS at 

P 

FIGURE 3-18 FIXED-STATION TURNSTYLE LOOP ANTENNA CONFIGURATION 

Entry I 

Coal Pillars 4 I Cross-Cut Alternative B 

Alternative A 

Source: Arthur D. Little, Inc. 

FIGURE 3-19 TURNSTYLE ANTENNA INSTALLATION POSSIBILITIES IN 

A COAL MlNE ENVIRONMENT 

Arthur D Little. Inc 
that creates an electrically rotating antenna and source field equivalent 

to that of a mechanically rotated one (one rotation per cycle of the 

excitation), thereby providing omnidirectional antenna coverage in the 

coal seam horizontal plane. 

Ideally, the two loops should have a common center as shown in 

Figure 3-18. This provides a common electrical phase center for the 

waves and minimizes mutual coupling between the two loops. Although it 

is possible to install two common-centered perpendicular vertical loops 

in coal mines by spanning the interesection of an entry and a cross-cut 

as in Figure 3-19a,it my not be a practical option in mine entries used 

by vehicles for the movement of men and material, unless special 

precautions are taken to protect the loop conductors. 

An alternative installation which avoids the need for such protection 

is shown in Figure 3-19b,whereby the loops are mounted on the perpendicular 

entry and cross-cut walls near the corner of a coal pillar. The noncoincident phase centers should not present any problems in the MF band 

of interest, but the mutual coupling between the loops may adversely 

affect the tuning and driving of the loops. The severity of these 

coupling effects, together with electrical and/or spatial means of 

minimizing any significant ones, will probably have to be eventually 

determined or verified experimentally. 

Finally, the most favorable length-to-height ratios, and number of 

turns, for fixed station rectangular loops will probably be best 

determined by experiment. The quasi-TEM mode of propagation excited in 

the coal seam, together with the thickness of the seam, are likely to 

impose limits on the increase in signal strength obtainable by increasing 

the length of a single-turn rectangular loop having a height approximately 

equal to the tunnel height in the coal seam. Once beyond the point of 

diminishing return with respect to loop length, additional increases in 

loop magnetic moment could be obtained by adding turns. Practical size 

constraints for typical in-mine installations could also lead to increased 

turns, as could the desire to increase the loop inductance to allow the 

loop to be resonated with capacitors that may be more convenientely 

available and practical. 

Arthur D Little Inc 
The wall-mounted loop presently used for the prototype MF mine 

wireless fixed installations in high-coal seams is a one-turn, 6 feet 

2 high by 24 feet long loop giving an NA = 12 m and a loop inductance 

of 27 ph. It is driven with a peak current of about 1 amp to produce 

2 

a magnetic xmemt of 14 amp-m . The bandwidth of 12 kHz requires a Q 

of 42 and a series-tuned real impedance level of about 2 ohms at a 

500 kHz operating frequency . Simple in-mine experiments should be performed to determine the 

relationship between loop length and signal strength produced in the 

plane of the loop at distances beyond about 100 - 200 meters from the 

loop. The results may reveal that a shape factor based on length-toheight ratio must be applied to the geometrical area to obtain an 

effective area from which the effective magnetic moment of the loop can 

be determined. A long piece of insulated wire capable of being driven 

from one end and any one of several other points along its periphery 

should be suitable for conveniently forming, mounting, and driving 

rectangular loops of different length on the wall of a mine tunnel. 

2. Alternative Mode Exciters 

a. Horizontal Grounded Lona Wires 

Mode excitation techniques or antennas (63964965966) similar to those 

used for the Navy's proposed worldwide ELF Sanguine communications system 

may also be applicable for mine wireless radio fixed installations. This 

section briefly describes some of the attractive similarities between the 

two applications, presents the results of some preliminary feasibility 

calculations for coal mines, and suggests some avenues of further investigation that appear promising. 

As discussed in Section 1I.D of this report, recent and continuing 

analyses (41'42943) and measurements (40,44) have shown that coal seams 

bounded above''and below by sedimenatary rock behave as lossy parallel 

plate waveguides with respect to the propagation of radio waves in the 

LF and MF frequency bands. At ELF,the conducting ionosphere and surface 

Arthur D Little Inc 
of the earth form the upper and lower boundaries of a similar parallel 

plate earth-ionosphere waveguide within which transmitted ELF waves travel 

around the world with very low attenuation, to be received by submerged 

submarines as depicted by Figure 3-20. Perhaps the most important difference betwen the two waveguides is that the coal seam waveguide is filled 

with a lossy medium, the coal itself, in addition to being bounded by 

lossy media above and below, thus introducing shunt loss in addition to 

the series loss of the "plates" in both waveguides. The spherical shell 

nature of the earth-ionosphere waveguide introduces only a different 

geometrical spreading loss factor. However, neither of these differences, 

the shunt loss nor the spherical shell, changes the basic nature of the 

quasi-TEM mode of propagation in the waveguide. In fact the Sanguine 

problem was initially treated in terms of cylindrically spreading waves 

in a f lat-earth , parallel plate waveguide. A spherical "focusing" 

correction factor was applied to the cylindrical wave solution afterwards. 

Thus, the excitation methods for one waveguide should be largely applicable to the other. 

The Sanguine investigators found that the most effective and feasible 

transmitting antenna configuration for their application was a parallel 

array of grounded, center-driven, long wire antennas. A single long 

horizontal wire is the simplest form of such an antenna. However, to 

achieve the required current moment IR requires an excessively long 

antenna (and thus land having the desired conductivity) and/or unreasonably 

high currents. High currents lead to two consequences that are undesirable 

for both the Sanguine and mine wireless applications, namely excessive 

power dissipation and environmentally nonpermissible field or current 

levels. A parallel array greatly reduces the maximum linear dimensions 

of the antenna, and distributes the required current moment over an area 

which in turn allows the use of several parallel wires, each carrying a 

fraction of the total current required. Furthermore, it was found that 

a spacing between horizontal wires (in the Sanguine case, 0000 AWG 

copper to carry 100 amperes) of about 2 skin depths provides a favorable 

compromise between overall system performance, power dissipation, and 

cost for the intended application. The reader is referred to references 

(63,64,65) for a more detailed treatment. 

Arthur D Little Inc 
Source: Arthur D. Little, Inc. 

FIGURE 3-21 SKETCH OF A GROUNDED LONG WIRE ANTENNA 

INSTALLED IN A COAL MINE TUNNEL 

Arthur D Little Inc 
Thus, one can visualize a mine wireless fixed station application 

that employs one or more horizontal long wire antennas strung along the 

roof of a mine tunnel and grounded to the rock above the coal seam by 

roof bolts as portrayed in Figure 3-21. The long wire antenna exhibits 

the same cos$ dipole pattern angular dependence as the vertical plane 

loop previously discussed. Thus, omnidirectional area coverage can be 

obtained in the same manner as with loops, by employing a second long 

wire antenna perpendicular to the first and driven 90' out of phase with 

it in turnstyle fashion as depicted in Figure 3-22. Note that the long 

wire antenna is better suited than a loop to turnstyle installations 

across the intersections of mine entries and crosscuts, because it does 

not require a floor-mounted wire which is generally difficult to protect 

from damage in well-travelled intersections. 

The most recent estimates (43) place the conductivity of roof rock 

in coal mines within the range of 0.01 to 1.0 mho/m with a value of about 

0.1 mho/m being typical. At an operating frequency of 300 kHz, this 

produces a skin depth in the rock, 6r, of about 2 meters. Thus, the 

Sanguine 26 separation criteria would allow at least two, and perhaps 

three, parallel long wires to be placed in a typical mine tunnel or entry, 

depending on the tunnel width in different mines or different parts of 

the same mine. Another possibility could involve a similar or greater 

number of considerably shorter long wire antennas installed in a crosscut tunnel between two entries as depicted in Figure 3-23. In each case, 

the individual grounded wires could be driven by separate transmitters 

keyed-on simultaneously via an audio twisted pair interconnection, or 

driven in-parallel by a single transmitter. 

The most effective configuration from the standpoint of power drain, 

communication range,ease of installation and maintenance, compatibility 

of fixed station units with the portable units, etc., would have to be 

determined after analyzing the results of a simple set of perf~rmance 

measurements in mines. Particular attention should be paid to the 

dependence of signal strength on the length of the long wire antenna 

(as suggested for the loop antenna measurements in Section III.D.~), and 

to the levels and variability of the impedance of simple roof bolt (and 

.111 

Arthur D Little. Inc 
I cos (wt - ~12) 

* I - I r. I cos wt 

Source: Arthur D. Little, Inc. 

FIGURE 3-22 LONG WIRE TURNSTYLE CONFIGURATION IN AN 

ENTRYICROSSCUT INTERSECTION 

Entry 

Crosscut 

Source: Arthur D. Little, Inc. 

FIGURE 3-23 LONG WlRE ARRAY CONFIGURATIONS IN A MINE ENTRY AND A 

CROSSCUT 

Arthur D Little Inc 
perhaps other types) ground terminations for the long wires. The 

dependence of both signal strength and antenna input impedance on the 

spacing between parallel long wires also needs to be examined in mines. 

The elaborate ground termination techniques available to a large 

and permanent, well-controlled Sanguine antenna installation will, of 

course, not be practical for a semi-permanent and largely uncontrolled 

mine installation. As a result, ground terminations achievable in a mine 

may be so large and/or variable as to make the horizontal long wire 

antenna impractical for mines. The spreading resistance of a 2 m long 

by 2.5 cm diameter ground rod (about the size of a roof bolt) making ideal 

contact along its length to rock of uniform conductivity 0.1 mho/m, is 

4.2 ohm,giving a total of 9.4 ohms for the two long wire terminations. 

Though 9.4 ohms is reasonable, it may not be achievable in practice. A - 

long wire antenna will also exhibit a nominal inductive reactance (63) 

which can be tuned out by a series capacitor if necessary. 

A first-order estimate of the performance of a single, terminated 

long wire compared to that of a conventional rectangular planar loop 

antenna can be made by computing the effective magnetic moment of the 

long wire and its return current path in the roof. When the antenna is 

long compared to the skin depth in the rock, the return current flow 

(63,64,65) spreads out into the roof to about one skin depth from the wire. 

Thus, the long wire and its return current path can be approximately 

represented by a rectangular loop of length R, effective height 6,/c, (65) 

and area A = R6 lJ2 . r 

An operating frequency of 300 kHz in rock having a conductivity 

= 1.0 mho/m, gives a 6r value of about 2 meters which results in an 

effective loop height of 1.4 m. The most effective long wire antenna 

length, a, may range from slightly longer than the waveguide (seam) 

height as in the Sanguine case, to a length that is on the order of the 

effective skin depth, 6, of the coal seam waveguide, where 6 is largely 

determined by the conductivity of the coal as opposed to that of the 

rock. The effective skin depth 6 of lossy coal seam waveguides at 300 kHz 

has been found (43) to vary from about 10 m in high loss seams, such as the 

Herrin No. 6 seam in Illinois, to as large as 100 m in low loss seams, 

Arthur D Little Inc. 
such as the Pittsburgh seam in West Virginia. Maximum wire lengths, Q, 

between 10 and 100 m can therefore be expected. Thus, the effective loop 

areas of grounded long wires are seen to compare favorably with that of 

the 2 m x 7 m rectangular fixed station loop discussed in Section III.D.l. 

Yet to be determined is whether the resultant waveguide coupling 

factor for the long wire is more or less favora-ble than indicated by the 

above simplified effective magnetic moment analysis, whether termination 

impedances comparableto the impedances of tuned loops can be achieved 

and easily maintained over time, and to what extent the signal strength 

remains proportional to wire length as the length is increased. Both 

analytical and experimental efforts will be required to resolve these 

issues. 

b. Vertical Grounded Wires 

The Sanguine ELF radio system unfortunately could not use what is 

perhaps the ideal antenna to excite the quasi-TEM mode in the earthionosphere waveguide, namely a vertical current source rising out of the 

conducting earth and terminating in the conducting ionosphere. Although 

such an antenna is clearly not feasible for the Sanguine application, it 

does become a very real possibility for use in the coal seam waveguide. 

In its simplest form it could consist of a vertical center-driven wire 

grmmded at each end to two vertical roof bolts, one in the roof and one 

in the bottom (floor), in a mine entry or a crosscut tunnel as depicted 

in Figure 3-24. It also has the advantage of producing omnidirectional 

coverage in the plane of the coal seam, without the addition of a second 

unft . 

Another perhaps more convenient configuration may also be feasible, 

namely the use of a steel, screw-tightened temporary roof support(Fig. 3-25) 

commonly used in the face area prior to permanent roof bolting. Such a 

roof support could be mated at each end with a special fixture designed 

to make a good grounding connection to the roof and bottom, and to 

provide a means for connecting the vertical rod coupler to the transmitter. This would avoid the need to drive a roof bolt or other type of 

grounding rod into the bottom. The shape of the fixture surface making 

contact with the coal would be designed with the objective of providing 

Arthur D Little Inc 
a. Grounded Vertical Wire b. Temporary Roof c. Temporary Support d. Temporary Support 

Configuration Support With Shaped Mated to Roof and 

Grounding Fixture Floor Bolts 

Source: Arthur D. Little, Inc. 

FIGURE 3-25 CONCEPTUAL SKETCHES OF VERTICAL WlRE 

MODE EXCITER CONFIGURATIONS 

- 

Arthur D Little lnc 
a favorably low terminat ion resistance, and a current spreading dis t ribution in tiie roof and bottom consistent with mode excitation constraints 

imposed by the skin depth in the rock. The shape of the contact surface 

must also lend itself to a convenient pressure-fit type of installation 

as the temporary support is screw-tightened against the roof and bottom. 

A similar result may also be obtainable without special shaping, by 

installing the vertical roof support so that it mates top-and-bottom to 

two previously installed roof bolt grounding rods. A secondary advantage 

of the structural-type of vertical mode exciter represented by the 

temporary roof support is that it allows the fixed station radio to be 

directly mounted to it, giving a physically integrated mine installation. 

A first-order estimate of the performance of this vertical wire/rod 

terminated source has been made and compared to that of a representative 

vertical plane loop. The vertical monopole-type source current generates 

a $-directed magnetic field component that is independent of $. Solution 

of the wave equation for this source distribution gives the following 

expression for the magnitude squared of H 4: 

for r>>6 where 6 is the effective skin depth in the coal seam waveguide, 

I is the source current, a is the attenuation constant, and B is the 

phase constant. Ideal matching of the source current distribution to 

that required in the rock by the mode is assumed. The corresponding 

field expression for a dipole source current such as that of a rectangular 

planar loop is 

for r>>6, where M is the source loop magnetic moment, 6 is the skin 

r 

depth in the rock, h is the height of the coal seam, and a and B are 

defined above. Forming the ratio of these two field quantities gives 

the expression 

Arthur D Little Inc 
practice. This coupling method, of course, will be subject to the same 

potential problems associated with establishing and maintaining good 

grounding connections to the rock as the horizontal long wire coupling 

method. Therefore, the likely success of both coupling methods will 

depend on the results of ground rod impedance experiments performed on 

roof bolts and other grounding devices implanted in the roofs and 

bottoms of mine tunnels. 

The conventional type of mine roof bolt installation, which contacts 

the rock at only two points, at the expansion wedge and at the base plate, 

will probably not provide a good enough electrical connection to the 

rock for the purpose of establishing and maintaining a low impedance 

ground. Some of the new plastic grouted bolts that fill the drill hole 

space between the steel roof bolt and the rock with a quick-hardening 

epoxy-type of plastic may create an even poorer connection if the 

grouting material is not conductive. However, if the plastic grouting 

material could be made conductive or of high dielectric constant, it 

- 

Arthur D Little Inc 
may be possible to approach the low terminal resistance of an ideal 

grounding rod, and to maintain it over a period of time sufficient to 

serve the semi-permanent needs of an advancing coal mine working section. 

At present, it appears that quantitative knowledge on how roof bolt and 

similar types of ground terminations work, vary over time, should be 

installed, and could be improved from an electrical standpoint, is not 

available. If and when this information becomes available, as a result 

of a carefully performed series of in-mine measurements, a more 

definitive assessment of the feasibility of the long horizontal wire 

and vertical rod mode exciting methods will be possible. 

3. Other Loop Antennas 

A considerable amount of development effort has also been devoted 

during the late 1960's and early 1970's to two other types of loop 

.antenna that are worthy of brief mention, namely self-resonant helical 

antennas by the University of Innsbruck (15,16,67,68,69) in Austria and 

large ferromagnetically loaded solenoidal loop antennas by Develco, Inc, (53) 

in the United States. Both efforts were aimed at generating very high 

2 magnetic moments (on the order of 1000 amp-m ) for special-purpose, fixedinstallation, through-the-earth, narrowband data communication applications 

in nonexplosive atmospheres, primarily in the VLF and LF bands. 

The Develco ferromagnetically loaded transmit loop antennas were 

designed for a bore hole through-the-earth data application subject to 

very high shock and overpressure. The antennas, consisting of nineteen 

ferrite rods surrounded by a long, high turns-density solenoid coil and 

a protective fiberglas shell, occupied a cylindrical volume about 10 ft. 

long and 1 ft. in diameter, and represented state-of-the-art electrical 

and mechanical designs optimized for the application. The University 

of Innsbruck transmit antennas were designed for through-the-earth data 

or code comunicat ion from an underground, environmentally controlled 

cavity, such as might be fashioned in an underground network of tunnels. 

The antennas were center-fed, self-resonant, air-core helical structures 

consisting of a large number of turns and typically occupying a volume 

on the order of a meter cube. Both development efforts also included 

analytical and experimental investigations of the associated through-theArthur D Little. Inc 
earth propagation problems, and methods of optimizing ferrite-loaded 

receive antennas. The Develco propagation effort was primarily concerned 

with vertical propagation through the overburden, and the University of 

Innsbruck effort was primarily concerned with horizontal propagation 

through an underground network of tunnels in a mountain. 

Although the specific antennas developed may be inappropriate for 

portable, intrinsically safe, mine wireless voice communication applications in the MF band, similar antennas and technology may in fact be 

applicable to permanently installed, one-way, wireless emergency warning 

or signalling applications in the ULF or VLF band for mines. The 

grounded horizontal long wire and vertical wire mode exciters discussed 

in the previous section may also be candidates for such applications. 

The high energy storage and/or high-power antenna/transmitter installations 

that would likely be required for long range, emergency warning applications could probably be located in areas of a mine where similar, highpower, nonintrinsically-safe equipment is allowed, or in speciallyprepa~dlocations where independent ventilation and sealing could be used 

to ensure a safe operating atmosphere even in the event of a mine 

disaster. Some of the antenna and signalling technology associated with 

the Develco ferrite-loaded cylindrical antenna systems may also be 

applicable to refuge shelter through-the-earth communications, especially 

in shelter situations where a large-area,air-core loop cannot be conveniently deployed. Further investigation will be required to determine the 

feasibility and suitability of these antenna and signalling techniques 

for narrowband emergency warning and refuge shelter signalling applications. 

Arthur D Little Im 
E. REFERENCES 

1. Chu, L. J., "Physical Limitations of Omni-directional Antennas," 

J. A. P., Vol. 19, pp 1163-1175, December 1948. 

2. Harrington, R. G., Time Harmonic Electromagnetic Fields, McGrawHill, New York, pp 264-316, 1961. 

3. Wheeler, Harold A. , "Fundamental Limitations of Small ~ntennas ," 

Proceedings of the I. R. E., pp 1479-1484, December 1947. 

4. Wheeler, Harold A., "Small Antennas," IEEE Trans. Antennas and 

Propagation, Vol. AP-23, No. 4, pp 462-469, July 1975. 

5. Collins, Robert E., and Zucker, Francis J., Antenna Theory, Part 1, 

McGraw-Hill Book Co., New York, 1969. 

6. Watt, A. D., VLF Radio Engineering, Pergammon Press, Oxford, 1967. 

7. Bohley, P., Walter, C. H., and Caldecott, R., "Development of the 

Multiturn Loop Antenna at H. F. for Shipboard ~~~lications," The 

Ohio State University Electro Science Laboratory, Final Report 

3518-1, Naval Ship Systems Command Contract N00024-73-C-1023, 

December 1973. 

8. Wheeler, H. A., h he Radiansphere Around a Small Antenna," Proc. 

IRE, pp 1325-1331, August 1959. 

9. Ikrath, K., Kennesbeck, W., and Hoverter, R. T., "~rees Performing 

as Radio Antennas," IEEE Trans. Antennas and Propagation, Vol. AP-23, 

No. 1, pp 137-140, January 1975. 

10. Hansen, Robert C., "Optimum Inductive Loading of Short Whip Antennas," 

IEEE Trans. on Vehicular Technology, Vol. VT-24, No. 2, pp 21-29, 

May 1975. , 

11. Jasik, Henry, Ed., Antenna Engineering Handbook, McGraw-Hill Book 

Co., New York, 1st Edition, 1961. 

12. Jinkings, P. W., "Antenna Systems for L. F. ~ransmitters ,'I Pointto-Point Communication, pp 32-45, January 1971. 

13. Seely, E. W. and Wiborg, P. H., Jr., "Horizontal VLF Transmitting 

Antennas Near the Earth," NOLC Report 721, Naval Ordnance Laboratory 

Corona, Corona, California, May 15, 1967. 

14. Collin, Robert E. and Zucker, Francis J., Antenna Theory, Part 2, 

McGraw-Hill Book Co., New York, 1969. 

Arthur D Little Inc 
Klein, Roy "Antennas for the VLF Region and LF Region," Scientific 

Report No. 4, AFCRL 70-0602, Contract No. F61052-69-C-0007, Univ. 

Doz. D. W. Bitterlich, VLF-Project, Univ. of Innsbruck, Innsbruck, 

Austria, September 1, 1970, NTIS No. AD719875. 

Grissemann, Ch., et al, "VLF/LF Antennas and Waves in Liquid and 

Solid Media," Final Scientific Report, AFCRL-71-0197, Contract No. 

F61052-69-C-0007, Univ. Doz. D. W. Bitterlich, VLF-Project, Univ. 

of Innsbruck, Innsbruck, Austria, December 31, 1970, NTIS No. 

AD725768. 

Kueckin, J. A. "Packset Radio Antenna Measurements," IEEE International Convention Record, Part 5, pp 261-270, 1966. 

Krupka, Zdenek, "The Effect of the Human Body on Radiation Properties 

of Small-Sized Communication Systems," IEEE Trans. on Antennas and 

Propagation, Vol. AP-16, No. 2, pp 154-163, March 1968. 

Shepard, N. H. and Chaney, W. G., "Personal Radio Antennas," IRE 

Trans. Vehicular Communications, Vol. VC-10, pp 23-31, April 1961. 

Maclean, T. S. M., and Morris, G. "Short Range Active Transmitting 

Antenna with Very Large Height Reduction," IEEE Trans. on Antennas 

and Propagation, pp 286-287, March 1975. 

Maclean, T. S. M. and Ramsdale, P. A. "Short Active Aerials for 

Transmission," Int. J. Electronics, Vol. 26, No. 2, pp 261-269, 

1974. 

Maclean, T. S. M. , and Ramsdale, P. A. , "Active Loop-Dipole Aerials," 

Proc. IEE, .Vol. 118, No. 12, pp 1698-1710, December 1971. 

Anderson, A. P., and Dawoud, M. M. "The Performance of Transistor 

Fed Monopoles in Active Antennas," IEEE Trans. on Antennas and 

Propagation, pp 371-374, May 1973. 

Fanson, Philip L. and Chen, Ku-Mu, "Instabilities and Resonances 

of Actively and Passively Loaded Antennas," IEEE Trans. on Antennas 

and Propagation, pp 344-347, March 1974. 

Dawoud, M. M., and Anderson, A. P., "Experimental Verification of 

the Reduced Frequency Dependence of Active Receiving Arrays," 

IEEE Trans. on Antennas and Propagation, pp 342-344, March 1974. 

111 

Arthur D Little. lnc. 
26. Daniel, Jean-Pierre and Terret,C., "Mutual Coupling Between 

Antennas--Optimization of Transistor Parameters in Active Antenna 

Design," IEEE Trans. on Antennas and Propagation, pp 513-516, 

July 1975. 

27. Lindenmeier, Heinz, "Active Receiving Antennas for Very Low 

Frequencies," Digest of the 1972 Group on Antennas and Propagation 

International Symposium,Williamsburg, Virginia, IEEE, New York, 

December 11-14, 1972. 

28. Turner, E. M., IEEE Conf. Rec., XVIII Annual Conference IEEE 

Vehicular Technology Group, 6-8 December 1967. 

29. Meinke, H., "Acktive Antennen," Nachrichtentech. Z., Vol. 19, 

pp 697-705, 1966. 

30. "~escription and Specifications for Underground Radio Transceivers 

for Use in the Underground Workings of Mines," Chamber of Mines of 

South Africa, Johannesburg, January 19, 1973. 

31. Vermeulen, D. J., and Blignaut, P. J., "Underground Radio Communication and Its Application for Use in Mine Emergencies," Trans. 

S. A. Institute of Electrical Engineers, Vol. 52, pp 94-109, 

April 1961. 

32. "Mystery Radio to the Rescue at Moorgate," New Scientist, p 641, 

March 13, 1975. 

33. "Fire Ground Communications Equipment, It Types PRD 2200/P~~ 2201, 

Product Brochure, The Plessey Company Limited, Hants, England. 

34. "Technical Specification for a Fire Ground Communications System," 

Plessey Types PRD 2200 and PRD 2201, The Plessey Company Limited, 

Hants, England, March 4, 1974. 

35. "'Inductorfone' Equipment," Mine Telephones and Signalling Equipment, Sheet 26, Plessey Communication Systems Ltd., Manchester, 

England. 

36. Dushac, H. M., "Two-Way Wireless Voice Communication System for 

Underground Coal Mines," Internal Report, Lee Engineering Division, 

Consolidation Coal Company, February 1976, and personal communications with Mr. Dushac. 

Arthur D Little Inc 
Chufo, R. L., Lagace, R. L., and Wilson L. R., "MF Mine Wireless 

Radio," Proceedings of Bureau of Mines Technology Transfer Seminar 

on Mine Communications, Pittsburgh, Pa., Bureau of Mines IC 

Washington, June 19 7 7. 

Bradburn, R. A. , "Communications for Haulage Loop Arounds , " 

Proceedings of Bureau of Mines Technology Transfer Seminar on Mine 

Communications, Pittsburgh, Pa., Bureau of Mines IC , Washington, 

June 1977. 

Bradburn, R. A., and Foulkes, J. D., "~ongwall Mining ~omrnunications," 

Proceedings of Bureau of Mines Technology Transfer Seminar on Mine 

Communications, Pittsburgh, Pa., Bureau of Mines IC , Washington, 

June 1977. 

Collins Radio Group, Rockwell International, "Mine Wireless Communication System," Final Report, Bureau of Mines Contract H0346067, to 

be published, Spring-Summer 1978. 

Lagace, R. L., Cohen, M. L., Emslie, A. G., and Spencer, R. H., 

"Propagat ion o f Radio Waves in Coal ~ines , I' Arthur D. Little, Inc . , 

Final Report, Task F--Task Order No. I., Bureau of Mines Contract 

No. H0346045, October 1975. 

Emslie, A. G., and Lagace, R. L., "Propagation of Low and Medium 

Frequency Radio Waves in a Coal Seam," Radio Science, Vol. 11, 

No. 4, pp 254-261, April 1976. 

Lagace, R. L., and Emslie, A. G., "Analysis of In-Mine Electromagnetic Wave Propagation Measurements," Arthur D. Little, Inc., 

Final Report, Task Order No. 4., Bureau of Mines Contract No. 

H0346045, to be published, Fall 1977. 

Collins Radio Group, Rockwell International, "~ropagation of EM 

Signals in Underground Mines," Final Report, Bureau of Mines 

Contract No. H0366028, to be published, May 1977. 

Zikun, G. A., Luzhnev, Yu. M., Pozhar, E. A., "Apparatus of Underground High-Frequency Mine Rescue Communication ' ~onetsk-1 M' , " 

All-Union Scientific-and-Research Institute of Mine Rescue Matters, 

Donetskiy Plant, U.S.S.R. 

Arthur D Little Inc 
Long, R. G. ,'I Acceptability Guidelines for Performance Characteristics of Underground Pager Phones," Arthur D. Little, Inc., 

Final Report, Bureau of Mines Contract No. 50166085, April 1977. 

Wolf, R. A. , "~esign of Electrical Equipment for Intrinsic 

Safety," Proceedings of the First WVU Conference on Coal Mine 

Electrotechnology, West Virginia University, August 2-4, 1972. 

Redding, R. J., INTRINSIC SAFETY--The safe use of electronics in 

hazardous locations, McGraw-Hill, London, 1971. 

Moon, P., and Spencer, D. E., Field Theory for Engineers, 

D. VanNostrand Company, Inc., Princeton, New Jersey, 1961. 

Wait, J. R., "Receiving Properties of a Wire Loop with a Spheroidal Core," Canadian Journal of Technology, Vol. 31, pp 9-14, 

January 1953. 

Dunlavy, J. H. Jr., "~esign Aspects of Ferrite Antennas in the 

Frequency Range Below 30 MHz," Proceedings of the 1968 Electronic 

Components Conference, Washington, D. C., May 9, 1968. 

Belrose, J. S. , "Ferromagnetic Loop Aerials ,I1 Wireless Engineer, 

Vol. 32, pp 41-46, February 1955. 

Rorden, L. H., Bacon, L. C., and Smith, R. L., "Underground 

Telemetry System Development--Design Definition phase," Develco, 

Inc., Interim Report on Defense Nuclear Agency Contract No. 

DNA 001-74-C-0298, July 1975. 

Snelling, E. C., "Ferrites for Linear Applications, Part I - 

Properties," IEEE Spectrum, Vol. 9, No. 1, pp 42-51, January 1972. 

Snelling, E. C., "Ferrites for Linear Applications, Part I1 - 

Performance Requirements ,I1 IEEE Spectrum, Vol. 9, No. 2, pp 26-32, 

February 1972. 

Sams, H. W., Handbook of Electronic Tables and Formulas, 2nd Ed., 

Herrington, D., and Meacham, S., Ed., Howard W. Sams and Co., Inc. 

Bobbs-Merrill Co., Inc., Indianapolis, 1962. 

Bohley, P., David, R. J., and Walter, C. H., a an-pack Loop 

Antenna System," The Ohio State University ElectroScience Laboratory, Final Report 3824-2, Naval Regional Procurement Office 

Contract N00123-74-C-0645, December 1974. 

Arthur D Little. Inc 
Davis, R. J., he Development of a Multiturn Loop Antenna for 

the AN/PRC-77 ,I1 The Ohio State University ElectroScience Laboratory, 

Technical Report 3824-1, Naval Regional Procurement Office Contract 

~00123-74-C-0645, December 1974. 

Bohley, P., and Newman, E. H., "Development of a VHF Multitum 

Loop Antenna for Seismic Sensor Systems," The Ohio State University ElectroScience Laboratory, Final Report Vol. ill, United States 

Army Materiel Command Contract DAAG39-72-C-0041, March 1973. 

Newman, E., Bohley, P., and Walter, C. H., "Two Methods for the 

Measurement of Antenna Efficiency," IEEE Trans. on Antennas and 

Propagation, Vol. AP-23, No. 4, pp 457-461, July 1975. 

King, H. E. , "Characteristics of Body-Mounted Antennas for Personal 

Radio Sets," IEEE Trans. on Antennas and Propagation, pp 242-244, 

March 1975. 

Reggia, F., "Low-Profile VHF Antennas for Seismic Detection Systems," 

HDL-TR-1635, U. S. Army Materiel Command, Harry Diamond Laboratories, 

Washington, July 1973, NTIS No. AD766703. 

Bernstein, S. I., et al, "~ong-Range Communications at Extremely 

Low Frequencies," Proceedings of the IEEE, Vol. 62, No. 3, pp 292- 

312, March 1974. 

Burrows, M. L. , "ELF Antennas, " Presentat ion at IEEE Antennas and 

Propagat ion Society Chapter Meeting, Washington, D. C. , October 19, 

1971. 

IEEE Transactions on Communications, Special Issue on Extremely 

Low Frequency (ELF) Communications, Vol. COM-22, April 1974. 

Wait, J. R., "Earth-Ionosphere Cavity Resonances and the Propagation of ELF Radio Waves," Radio Science, Vol. 69D, pp 1057-1070, 

August 1965. 

Nessler, N., "Theoretical and Experimental Studies of VLF and LF 

Waves," Final Scientific Report--1973, AFc~~-T~-73-0174, Contract 

No. F 44620-72-C-0052, University of Innsbruck, Innsbruck, Austria, 

February 28, 1973, NTIS No. AD753778. 

Nessler, N., "Considerations on the Dimensioning of Ferrite 

Antenna Amplifiers in the VLF Regions," Scientific Report No. 2, 

Arthur D Little Inc 
AFCRL-69-0420, Contract No. F61052-69-C-0007, Univ. Doz., Dr. W. 

Bitterlich, VLF-Project, University of Innsbruck, Innsbruck, Austria, 

July 15, 1969, NTIS No. AD695489. 

69. Lukavec, R., "~easured Properties of Small Self-Resonant Helical 

Antennas at Low Frequencies for Application in Dissipative Media, " 

Scientific Report No. 8, AFCRL-69-0096, Contract No. 61(052)-902, 

Univ. Doz., Dr. W. Bitterlich, VLF-Project, University of Innsbruck, 

Innsbruck, Austria, December 1, 1968, NTIS No. AD684578. 

Arthur D Little Inc 
(This page intentionally left blank) 

U 

Arthur D Little. Inc 
IV. COUPLING TO EXISTING CONDUCTORS IN MINES 

A. SUMMARY 

To evaluate the potential advantages and practicality of extending 

mine wireless ranges by coupling to existing conducting structures in 

mines, theoretica1.equations were developed,relating loop antenna strength, 

frequency, and its distance from two-wire cables and structures, to the 

current and voltage induced in that structure by the loop antenna. These 

equations allow one to compute not only the loop-to-structure coupling 

but also loop-to-loop coupling via the two-wire structure. Results computed for three frequencies (3, 100, and 1000 kHz) indicate that: 

a performance improves with closeness to the two-wire structure, 

increased separation of the wires,and increasing frequency up 

to about 0.5-1 MHz in a mine tunnel; 

a loop-to-loop communications along a haulageway appear practical 

in the presence of a trolley wire system and may also be 

usable in the presence of untwisted cables and single wires 

attached to the ribs; 

a coupling to twisted pair cables is severely reduced over that 

for untwisted pairs ; 

a performance should be significantly reduced for loops located 

in tunnels adjacent to the one in which the conductors are 

located. 

The results for representative cases of interest are plotted. 

B. INTRODUCTION 

The objective is to calculate the coupling between a loop antenna 

which can reasonably be carried by a miner and conductors which are commonly found in mines. The antenna loop has a magnetic moment of M=NIA 

where N is the number of turns, I is the current in the loop, and A is 

its area. For an antenna which is wrapped around a man, a reasonable 

2 

source strength is M= 112 to 4 amp-turn meter . We will use the value 

1 amp-turn meter2, here. The conductors available for coupling to are: 

a telephone lines (twisted and untwisted pairs); 

trailing cables (either twisted pairs, twin-leads or more 

complex structures); 

a trolley wirelrail (approximated here by a two-wire line) 

and 

power distribution lines(a conventional 3-phase cable for mine use). 

Arthur D Little. lnc 
The simpler, in-air case will be examined to get upper bounds for performance, These results should be applicable within the haulageways where 

the conductors are located, Transmitter requirements will be more severe 

for coupling beyond the immediate tunnel cross-section in which the conductors are located because of the additional losses introduced by the 

coal and surrounding rock, 

C , REPRESENTATIVE GEOMETRIES 

We first examine several different cases of coupling between a 

loop, assumed of infinitesimal size, and twin-lead and twisted-pair 

conductors, The assumption of an infinitesimal loop is justified if 

the diameter of the loop, approximately 0.3 m, is much less than the 

wavelength of the signal, and the loop-to-conductor spacing, Table 4-1 

summarizes several situations of interest, Figure 4-1 shows several basic 

arrangements of the transmitting loop with respect to simple twin-lead 

conductors, If the miner wears the transmitting antenna as a bandolier 

around his body, then situationsla andlb represent cases where the 

planes of the twin-leads and the loop are horizontal; while situations 

lcand Idrepresent cases where the plane of the twin-lead cable is vertical but the loop itself is still in the horizontal plane, Situations 

leand lf represent cases where the cable direction is parallel to the 

TABLE 4-1 

TYPICAL DIMENSIONS FOR STRUCTURES 

OF INTEREST IN A MINE 

Frequencies 3kHz to lMHz 

Wavelengths 10 5 m to 3xlOm 2 

Trolley Lines 

(trolley wire to lm (low coal) to 2m (high coal) 

track spacings) 

Twisted Pairs 

(cable diameters) O.O2m to 0.07m 

Flat Twin-Lead 

(lead spacings) 0.015m to 0.05m 

Three Conductor Cables 

(cable diameters) 0.02m to 0.08m 

Antenna-to-Cable 

Distances 2m 

Source: Arthur D. Little, Inc., Essex and Anaconda Catalogs 

Arthur D Little Inc 
la. I,$//

Coplanar Horizontal Arrangement; Cable. 

I 

I I 

I 

& I 

I 

Source: Arthur D. Little, Inc. 

Aligned Arrangement; 

Horizontal Cable. 

Edge-on Arrangement; 

Vertical Cable. 

A1 igned Arrangement; 

Vertical Cable. 

Edge-on Arrangement; 

Cable Direction Parallel to Loop Ax is. 

90' Edge-on Arrangement; 

Cable Direction Parallel to Loop AXIS. 

FIGURE 4-1 DIFFERENT GEOMETRIES FOR A HORIZONTAL 

LOOP ANTENNA AND A TWIN-LEAD CABLE 

Arthur D Little Inc 
axis of the loop, but in case E the plane of the twin-lead cable contains the loop axis, while in case F the plane of the twin-lead cable is 

perpendicular to a plane containing the axis of the loop. These latter 

two situations where the direction of the cable is parallel to the axis 

of the loop are not considered as realistic alternatives. Figure 4-2 

shows similar arrangements for a twisted-pair but since there is no 

unique plane which contains only the two wires of the twisted-pair, 

there are less cases of interest. 

In general, the transmit loop antenna and the mine structure that 

is acting as the conductor will not have such simple relationships as 

those shown in Figures 4-1 and 4-2. The coplanar relationships in la and 

lb are reasonable approximations for most situations where a horizontal 

twin-lead or a twisted-pair are mounted on the side wall of a tunnel 

or hung from the roof. In low coal, the conductor is unlikely to be 

more than 112 m above or below the plane of the transmitting antenna 

which may be 2 m to 600 m away. At 2 m the angle above or below the 

transmitter loop plane is 14O while at 600 m the angle is 0.05O. 

Cos 14O is about 0.97, so the field strength at a conductor 112 m 

above the plane of the transmit loop is approximately equal to that for 

a conductor in the plane of the transmit loop. Assuming a 2 m high 

seam for high-coal, the maximum angle of the conductor above the plane 

of the transmitter loop is 37O, which has a cosine of 0.80, so the 

absence of coplanarity is more significant here. However, at distances 

greater than 6 m, the coplanarity assumption is a reasonable approximation, as shown in Figure 4-3. 

Simil-arly, when the twin-lead cable is in the horizontal plane and 

suspended from the roof away from a wall, case la is still a reasonable 

description of the situation as the transmitting loop approaches from 

600 m to 6 m in high-coal, and to 2 m in low-coal. There can then be a 

transition region as the transmitting loop moves under the cable until 

the situation of lb is achieved. 

.I. 

Arthur D Little. lnc 
Coplanar Arrangment 

(similar to la and Ic) 

Aligned Arrangement 

(similar to 1b and Id) 

Edge-on Arrangement 

(similar to le and If) 

Source: Arthur D. Little, Inc. 

FIGURE 4-2 DIFFERENT GEOMETRIES FOR A HORIZONTAL LOOP 

ANTENNA AND A TWISTED-PAIR CABLE 

99 Arthur D ~ittle lnc I 
Good to 3% Over This Range 

Horizontal 

Center Line of Twin Lead 

he Assumption of Coplanarity is Poor Separation 

Aligned Arrangement Valid Over This Range on a Log Scale (m) 

Source: Arthur D. Little, Inc. 

FIGURE 4--3 RANGES OF VALIDITY OF COPLANARITY ASSUMPTIONS 

- 

Arthur D Lit tlo Inc 
Similar ranges of validity apply to the case where the twin-lead 

cable is in the vertical plane, A special case of this is the trolleywire system, which because of its size with respect to the tunnel dimensions, the loop position will always lay vertically somewhere between 

the trolley-wire and the rail return, 

D, COUPLING TO SIMPLE TWO-WIRE LINES 

1. Case la - Coplanar Geometry 

a, Electric Field Method 

This case has been considered in detail by Albert A, Smith, Jr, in 

a paper called "The Response of a Two-Wire Transmission Line Excited by 

the Non-uniform Electromagnetic Fields of a Nearby Loop, " published in 

the IEEE Transactions on Electromagnetic Compatibility, EMC-16, 196-200 

(1974). The equations given in this paper for the near, or inductivefield, case are valid over most of the ranges of parameters of interest 

to us, see Table 4-2. The variable BR = 2mR/X should be much less than 

unity; R is the spacing between the loop and the cable and X is the 

wavelength of the signal in air. 

TABLE 4-2 

REGIONS OF VALIDITY FOR USING BR <<I CRITERION, 

THE INDUCTIVE-FIELD APPROXIMATION 

(IN AIR) 

f (kHz) 3 100 1000 

The entries in the table are values of BR = ~ITR/X in air. 

(1) rans sit ion zone between inductive and radiation fields. 

(2) Radiation - field zone. 

Source: Arthur D. Little, Inc. 

Arthur D Little Inc 
If the two-wire line is represented as a transmission line terminated at both ends by impedances, Smith has developed several simplified 

expressions for the current induced in the load resistance terminating 

the transmission line. Figure 4-4 defines the quantities of interest in 

the expressions. The loop is at a distance R from the center line of 

the transmission line and at a distance w from the start of the line. 

The transmission line is s long and consists of two parallel conductors 

spaced b apart. Both the loop and the transmission line are in the 

plane x-z. The electric field in the plane x-z due to the loop is a 

function of r only. In most mine situations, s is a large dimension, 

at least several hundred meters, b << R (see Table 4-1 for values of b), 

and the loop is not close to either end of the transmission line. 

Then, according to Smith, the following approximations can be made: 

a EZ = a constant depending on R over a length of the 

transmission line L = R. 

a EZ = 0 over the rest of the line. 

a E = a constant depending on R over a length of the 

X 

transmission line L = R, and 0 over the rest of the line. 

A current is induced in the transmission line because of the non-uniform 

distribution of these fields. In the case of a radiation field, the 

current is principally induced by the phase difference of the wave over 

the two conductors. 

The normalized expression for the current in the load at the end 

of the line is given by: 

where 2 sin (PL/2) 

E (R, 4 DP 

(1) From Smith IEEE article previously cited. 

102 

Arthur D Little Inc 
Source: Smith, IEEE Trans EMC, EMC-76 196-200 (1974) 

FIGURE 4-4 GEOMETRY OF THE TRANSMITTING LOOP AND TRANSMISSION LINE 

FOR THE CASE OF FIGURE 4-l(a) 

Arthur D Little Inc 
I2 (s94 bZo R2 w b(Zo cosps + jZ1 sin 0s) R2 (S - W) (1) 

and -------- - E(R,w) D[R~+W~]~'~ Q1 + D [R2 + (s - w)~] 3'2 Q2 (3) 

1 + (b/2R)2 - j(b/R) cot (pbl2) ('1 

p=- - 

[l - (b/2R)'] 9 

D = (ZoZl+ZoZ2)cos/3s + j(z; + ZlZ2) sines 

/3 = 27r/h phase constant 

h = wavelength 

Zo = 276 loglo (200b/a) characteristic impedance (1) 

b = spacing (meters) 

a = diameter (cm). 

Since s is usually larger than R and the loop is not near the ends of 

the line, it is reasonable to place E = 0 as well as EZ = 0 at the 

X 

ends of the line. Then, only the term Il(s,w)/E(R,~ due to E z over 

the length of line L = R near the loop contributes to the current in the 

line. 

- 

Arthur D Little. Inc. 
As R grows larger, the nature of the loop-caused field at the line 

changes from inductive to radiative. Then, the equations are modified 

to: 

except 

(11 I1 (s, a) - . . . as above . . . 

E (R, a) 

1 - j(b/2R) cot (pb/2) ('1 

P = 1 - (b/2~)~ 9 

I2 (s, a) bZo Rw b(Zo cosps + jZ1 sinps) R(s -w) (1 1 

and - 

E (R, a) D(R? + w2) QI + D [R~ + (S - w)~] Q2* (12) 

with Q1 and Q2 as above and Z = Z2 = ZO. 1 

These complex expressions must be rearranged so that their magnitudes can be evaluated. Since the inductive-field assumption is valid 

over most of the ranges of interest to us, and since it is reasonable 

to assume that w > R for all cases of interest to us, only the expression for I is rearranged. Then we have: 1 

sin (+) sin 

If R << A, and b << R, then; 

Arthur D Little Inc 
The value of this transfer admittance 

has been evaluated below for three cases: 

1. Twin-lead, telephone and power cable 

b = 0.04 m; Zo = 280.6 ohms 

R - < 20 m; T = -77dB 

R - > 200 m; T = -78dB 

* 2. Low coal, trolley system (Two-wire line apnroxi~ation) 

b = 1 m; a = 0.05 m; Z = 442.4 ohms 

0 

R - < 20 m; T = -53dB 

R - > 200 m; T = -55dB 

* 3. High coal, trolley system (Two-wire line approximation) 

b = 2 m; a = 0.05 m; Z = 525.3 ohms 

0 

R - < 20 m; T = -48dB 

R - > 200 m; T = -50dB 

Table 4-3 lists the values of electrical field strength in the plane of 

the loop at 2 m from a loop antenna in air as a function of frequency. 

It also indicates the reduction in field strength at increasing distances. 

Using these field strengths and the values of T given above, the values 

of current induced in the conductors and the matched loads have been 

calculated and tabulated in Table 4-4. 

b. Magnetic Field Method 

The value of these induced currents can also be estimated by an 

equivalent simpler method based on the voltage induced in the pair by 

the loop magnetic flux cutting the pair. In the above coplanar case, 

the 0-component of the loop field, He, would be used, and the induced 

voltage estimated by using the approximation of an equivalent constant 

flux density distribution B =p H (R) linking an area enclosed by the 8 08 

separation b of the cable pair over a length L along -the pair. Here, L 

is equal to ZR, instead of R, where R is the shortest distance from the 

center of the loop to the midpoint between the cable conductors as before, 

and H (R) is the value of loop magnetic field at this midpoint. 8 

* In practice, the trolley line characteristic impedance is closer to 

200 ohms, because of the large effective diameters of the trolley 

wirelfeed conductors and the rail return conductors. 

I Arthur D Little, Inc. 
TABLE 4-3 

VALUES OF E IN THE PLANE 9 

OF A LOOP AT A DISTANCE R AWAY IN AIR 

2 (Source Strength M = 1 amp-meter ) 

Frequency 3kHz lOOkHz lMHz 

R(m) 

2 480pv/m 15,80Opv/m 158,0OOpV/m * * * 

20 -40dB -40dB -39dB 

200 -80dB -79dB -67dB 

500 -96dB -93dB -76dB 

Source: Arthur D. Little, Inc. 

*dB reduction with increasing distance relative to 2 meter values 

Arthur D Little Inc. 
TABLE 4-4 

CURRENTS IN p AMPERES TNDUCED IN CONDUCTORS 

FOR THE SITUATION OF FIGURE 4-la 

(IN AIR) 

(a) Frequency = 3kHz 

b (m) 0.04 

R (m) 

2 0.06 

(b) Frequency = lOOkHz 

(c) Frequency = lMHz 

Source: Arthur D. Little, Inc. 

Entries in the Table are currents in PA for a source strength 

of M = 1 amp-meter2 and matched terminations Z = Z2 = Z0 1 

* Dash indicates that inductive field approximations do not 

apply at these ranges for the indicated frequencies. 

I 

Arthur D Little Inc. 
In the inductive field region of interest, examination of the 

reciprocal problem of coupling from a long two-wire line to a small 

loop reveals that the mutual coupling impedance Zm = Vloop/Iline, is 

identical for both the coplanar-la and coaxial (aligned) - lb configurations. Therefore, for the aligned case (lb) one can simply use the 

coplanar values computed above, or utilize a value of L equal to R, (as 

used in Smith's approximate E-field method) in order to compensate for 

the factor of two introduced by the Hr field that now links the cable, 

and which is twice the strength of the % loop field in the inductive 

field region. The approximate flux linkage method of computation is 

illustrated below for this coaxial (aligned) case lb in which the 

r-component, Hr, of the loop field is the appropriate one to use. 

(This result has been verified by evaluating the appropriate integrals.) 

2. Case lb - Aligned Geometry, Magnetic Field Method 

To estimate the voltage induced in the twin lead cable conductors 

in the aligned case lb, the H field strength will be assumed constant 

r 

over a line length L = R, and zero elsewhere on the line, for computing 

the effective flux linking the cable as noted above. Therefore the 

required H field strength is that produced by the loop at a distance 

r 

R, which again is the shortest distance from the center of the loop to 

the midpoint between the cable conductors. The effective area being 

linked by the flux is given by Lb = Rb. The relevant expressions are: 

where the symbols have their usual meaning. Now the flux is given by: 

$ = Ba - po H, (Rb) (18) 

Arthur D Little Inc. 
and by ~enz's law, the induced series voltage is given by: 

Substituting the expression for H and simplifying, leads to the 

r 

expression: 

If 27rR << A, then 

The corresponding expressions for the coplanar case - la, which 

utilize the loop He field component at 8= n/2 are: 

V " w PO (2Rb) Hg 

and 

which reduces to Equation (21) when 2nR<<X, as it should. 

-. 

Arthur D Little, Inc. 
Table 4-5 presents the results of these calculations for M = 1 amperemeter2 and the parameters listed. The corresponding currents induced 

in the load at each end of the matched cable is then given by: 

The equivalent transmission line excitation and terminations are shown 

in Figure 4-5. As discussed above, the induced voltage is identical for 

both the aligned and coplanar (shown) geometries . The above technique 

can also be applied to the cases of arbitrary loads Z and Z2 terminating 1 

the transmission line, 

Loop Transmitter 

Source: Arthur D. Little, Inc. 

FIGURE 4-5 EQUIVALENT TRANSMISSION LINE CIRCUIT FOR INDUCED 

VOLTAGE AND CURRENT (Case of Z1 = 22 = ZO) 

3, Unfavorable Geometries 

Because the plane of the cable conductors is rotated through 90' 

between the situations of Figures 4-la and lc and Figures 4-lb and Id, 

the flux linkage in situations approximating those of Figures 4-lc and Id 

is substantially reduced below those of the two cases considered above, 

causing a corresponding major reduction in the magnitude of the induced 

signals. Alternatively stated, in Figure 4-lc, the values of E at 0 

the two conductors are equal because they are the same distance from 

Arthur D Little fnc. 
TABLE 4-5 

VOLTAGES INDUCED IN CABLE CONDUCTORS 

IN THE SITUATION OF FIGURE 4-lb 

A=) 

(a) Frequency = 3kHz 

b (m) 0.04 1 

R (m) 

2 37 . 6pV 940pV 

20 0.38pV 9.6pV 

200 3.8nV 9 6nV 

500 0.6nV 15nV 

(b) Frequency = lOOkHz 

b (m) 0.04 

R (m) 

(c) Frequency = 1MHz 

b (m) 0.04 1 2 

R(m) 

2 12.6mV 0.32V 0.64V 

20 136pV 3.4mV 7 .OmV * -------------------------------- 

200 5 . 4pV 138pV 276pV 

500 2.2pV 54pV 108pV 

Source: Arthur D. Little, Inc. 

Voltage values are for a source strength of M = 1 amp-meter2 and 

matched terminations Z1 = Z2 = ZO. 

* Values of voltage above the dashed lines can be compared with the 

corresponding current values in Table 4-4 by dividing the above 

voltages by 22 . Values of voltage below the dashed line are 

coarser estimaees for the transit ion and radiative field regions. 

LIX Arthur D Little, lnc. 
the loop, while in Figure 4-ld the E-field at the conductor is effectively zero. In both cases no lines of magnetic flux link the conductors. 

Similarly, the signals induced by the loop antenna in arrangements approximating Figures 4-le and 4-lf are also significantly less 

than those of cases la and lb because the net flux linkages are zero 

as shown, and still insignificant for small deviations from the alignments shown. . 

E. COUPLING TO TWISTRI-PAIR CABLE 

We have not performed similar calculations for the twisted-pair 

cases because the paper "Predicting the Magnetic Fields from a TwistedPair Cable" by J. R. Moser and R. F. Spencer, IEEE Trans. EMC, EMC-10, 

324-329 (1968) states that the field from a twisted pair is reduced by 

50dB from the field from a twin-lead at a distance d = 1.5p from the 

cable, where p is the pitch of the twisted pair, assuming both wires 

are carrying the same current. Assuming reciprocity, this implies a 

reduction of at least 50dB in the inward coupling parameter for all 

twisted-pairs with p <1 m, which is still a conservative relatively 

large (or loose) pitch in practice. 

F. LOOP-TO-LOOP COUPLING VIA THE LINE 

By the principle of reciprocity it is possible to use the loop-toline coupling equations to compute the coupling between a transmit and 

a receive loop via the Line. The equations developed above in D.2 for 

the induced voltage in the line using the magnetic flux linkage approach 

are particularly convenient in this regard, although the equations using 

the E-field approach to compute the induced line current can also be 

utilized by first dividing by 2Z0 to get the voltage. 

Proceeding with the induced voltage approach in D.2 we have, by 

reciprocity, 

Arthur D Little Inc 
where Z is the mutual coupling impedance, and V is given by m oc-line 

Equations (20), (21) or (23), with M in these equations given by 

Therefore, 

so line-to-loop coupling for a given value of I can be computed by line 

substituting (20). (21) or (23) and (26) with NA = NrA into (27), 

r 

where the subscript r denotes receive loop; and loop-to-loop coupling 

via the line can be computed by further substituting (24) for Iline 

and (26) with NA = N A into (27), where the subscript t denotes 

t t 

transmit loop. 

When the ranges from the loops to the line are much smaller than 

the wavelengths, a commonly encountered practical case in coal mines, 

Equation (21) can be used for Voc line - Then for line-to-loop coupling . (27) becomes 

and for loop-to-loop coupling (27) becomes 

Equations (28) and (29) are the formulas which apply for a matched 

lossless transmission line. For a lossy matched line, (28) and (29) 

would be multiplied by e-aR where cc is the line attenuation in nepers 

per meter, R in (28) is the distance in meters along the line from the 

signal generator to the loop, and R in (29) is the length of line in 

meters separating the transmit and receive loops. Equations (28) and 

(29) can be used to calculate numerical results for the coupling 

factors between line and loop and loop-to-loop via the line, and the 

absolute signal voltage levels induced in the receive loop, for typical 

guided wireless configurations in mine haulageways. 

C 

Arthur D Little Inc 
G. SIGNAL-TO-NOISE RATIO ESTIMATES 

Estimates of output signal-to-noise ratios can be obtained by 

comparing the signal levels given by (28) and (29) with the corresponding 

output noise voltages induced in the receive loop by the ambient H-field 

noise, or by comparing the measured H-field noise levels in the air with 

the calculated H-field signal levels in the air given by equations (28) 

and (29) divided by 2nfpNrAr. Namely, for line-to-loop communications, 

the received H-field signal level is 

and for loop-to-loop communications, the received H-field signal level is 

Equations (28) and (30) are convenient to use in cases where the 

line is direct-driven by a signal source, as in the case of a trolley 

wire carrier system; whereas (29) and (31) apply to cases where a matched 

dedicated line or trolley wire is inductively driven by a loop-fed signal 

source. 

Figures 4-6 and 4-7 compare received H-field mine noise levels and 

calculated signal levels for a loop-to-loop via the line communications 

link. The curves are plotted versus frequency. A narrow-band FM radio 

communication system having an IF bandwidth B = 12 kHz, receiver noise 

figure F = 6 dB, transmit air-core loop magnetic moment M = I N A = 

2 t t t t2 

1 amp-m , and receive air-core loop effective turns area N A = 1 m , rr 

similar to a system presently being considered for wireless mine communication applications by Collins Radio Group on Bureau of Mines Contract 

H0346047. The rms noise levels chosen to characterize the mine electromagnetic noise environment were derived from the sample of magnetic 

Arthur D Little Inc 
RMS Magnetic 

Field Strength 

in dB re 1 pamp 

per meter 

Legend for Mine Noise Curves Shown on Figure 2-7 

Noise Levels Based on a 12 kHz Receiver Bandwidth 

Two-Wire Line Conductor Separation - b = 2 Meters 

Transmit Moment - Mt = 1 amp - m2 

Signal Attenuation Along Transmission Line Not Included 

Frequency in kHz 

Source: National Bureau of Standards (Report NBSI R 74-389, June 1974) and Arthur D. Little

FlGURE4-6 COMPARlSONOFLOOP-TO-LOOP-VIA-THE-LINESlGNALLEVELSINAIR 

WITH EQUIVALENT RECEIVER NOISE AND REPRESENTATIVE RMS MAGN

FIELD NOISE LEVELS MEASURED IN THE McELROY COAL MINE 
Legend for Mine Noise Curves Shown on Figure 2-8 

RMS Magnetic 

Field Strength 

in dB re 1 pamp 

per meter 

Noise Levels Based on a 12 kHz Receiver Bandwidth 

Two - Wire Line Conductor Separation - b = 2 Meters 

Transmit Moment - Mt = 1 amp - m2 

Signal Attenuation Along Transmission Line Not Included 

Frequency in kHz 

Source: National Bureau of Standards (Reports NBS Technical Note 654, April 1974 Robena; 

and NBSlR 74-390. Jt~ne 1974, Itmann) and Arthur D. Little, Inc. 

FIGURE 4-7 COMPARISON OF LOOP-TO-LOOP-VIA-THE-LINE SIGNAL LEVELS IN AIR 

WITH EQUIVALENT RECEIVER NOISE AND REPRESENTATIVE RMS MAGNETIC FIELD 

NOISE LEVELS MEASURED IN THE ROBENA AND ITMANN COAL MINES 
field, time-averaged, rms noise levels measured (with 1 kHz bandwidth 

instrumentation) at spot frequencies in the frequency range of interest 

in three coal mines by Bensema, Kanda, and Adams of the National Bureau 

of Standards. The measurement conditions and locations for these noise 

levels are the same as those described in the comprehensive legends of 

Figures 2-8 and 2-9. Also shown is a curve representing equivalent 

field strength for the intrinsic receiver noise. 

In Figures 4-6 and 4-7, signal levels are shown for two transmitter 

and receiver configurations of interest relative to the location of a 

matched trolley wirelrail or dedicated wirelrail transmission line in a 

high-coal haulageway tunnel. The transmission line conductor separation, 

b, is 2 meters and the line characteristic impedance,Zo , is 300 ~hms. 

The first configuration is for the case when both transmit and receive 

units are located at different stations along the haulageway, but at 

the same distance of 2 meters from the transmission line and with the 

planes of both loops parallel to the plane of the transmission line. 

The second is for the case when only one unit is located 2 meters from 

the transmission line and the second unit is located 20 meters away from 

the line, perhaps in an adjacent entry. These signal curves apply 

strictly for an air medium, which the 2 meter case in a mine tunnel 

closely approximates. However, only minor downward corrections are 

necessary (and only at the highest, most favorable coupling frequencies) 

to apply the 20 meter results to the case of a homogeneous, moderately 

conducting medium ( a= ~ho/m). This can be seen by comparing the 

"30 meter" in-air and conducting medium curves in Figures 2-8 and 2-9. 

Figures 4-6 and 4-7 show that the mine noise levels generally 

diminish with increasing frequency (approximately as l/f), and that 

mine generated noise greatly exceeds receiver noise under most mine 

operational conditions in haulageways and near section mining machinery 

and power conversion equipment. These figures also show that signals 

which have been coupled onto and off transmission lines, such as trolley 

wirelrails, are high when the portable wireless transmitlreceive units 

are close to these lines (the Rr = Rr =2 meter case); and that the 

signal field strength at the receiver increases proportional to frequency 

if the signal attenuation caused by resistive losses along the cable is 

neglected. 

118 

- Arthur D Little Inc 
A reasonable radio circuit grade of performance for a wireless mine 

communication system is the Circuit Merit Figure #3 classification 

(occasional message repetitions required). This is the minimum figure 

normally considered for commercial radio service. It requires that the 

average rms carrier-to-noise ratio at the receiver be at least 10 dB 

or better for the mine noise conditions prevailing during the communication intervals. Comparing signal and noise levels according this 

criterion, we find that loop-to-loop communications via the line can be 

established over significantly longer ranges than those obtainable - in 

the absence of the line (Chapter 11) 

Figures 4-6 and 4-7 also suggest that, the higher the frequency 

above about 100 kHz, the better performance will be. However, in practice, 

the attenuation suffered by the signal while travelling along the transmission line (due mainly to resistive losses in the tunnel walls) also 

increases with frequency. This attenuation loss will cause the plotted 

signal curves to eventually droop downwards as the frequency is increased, 

with the number of dB to be subtracted from the signal curves depending 

on the length of transmission line between the two portable units. Theoretical analyses conducted by J. Wait* of NOAA indicate that the attenuation rate along unloaded trolley wire/rail transmission lines in mine 

tunnels can vary from about 1 d~/km at 100 kHz to about 10 d~/lcm at 

1 MHz. Wait** has also shown that similar behavior, but at slightly 

higher attenuation rates, can be expected with single power cables 

* J.R. Wait and D.A. Hill, "Radio Frequency Transmission Via a Trolley 

Wire in a Tunnel with a Rail Return", to be published in the March, . 

1977 issue of IEEE Transactions on Antennas and Propagation 

D.A. Hill and J.R. Wait, "Analysis of Radio Frequency Transmission 

Along a Trolley Wire in a Mine Tunnel", IEEE Transactions on Electromagnetic Compatibility, November, 1976. 

**J. R. Wait and D.A. Hill, "Low-Frequency Radio Transmission in a 

Circular Tunnel Containing a Wire Conductor Near the Wall", 

Electronics Letters, 24 June 1976, Vol. 12, No. 13, pp. 346-347. 

Arthur D Little Inc. 
arid phone lines hanging from the roof of a mine tunnel (in this case the 

return current path is in the surrounding rock). The method of images 

can also be a useful tool for gaining understanding of signal behavior 

for such configurations. Taking these attenuation rates into consideration for a 1 km length of transmission line, the signal level curves 

can be expected to start decreasing above about 500 kHz, thereby defining 

a somewhat broad optimum operating region beween approximately 200 kHz 

and 1 MHz. Recent tests conducted by Dushac* with Plessey Telecommunications SSB portable radios operating at 425 kHz along trolley haulageways in several coal mines have shown that loop-to-loop via the line 

communication ranges in excess of 112 to 1 km are attainable with practically-sized portable units, and that such units are well suited for 

use in haulage loop-around operations for communication between the 

locomotive operator and the snapper who couples and uncouples coal cars. 

When one of the portables is located in an adjacent entry (the 

Rr = 2m, R = 20 m case) signal levels will be substantially reduced t 

(by about 40 dB). However, one can also expect lower noise levels (by 

about 20 dB), thereby effectively decreasing the noise curves relative 

to the Rr = 2m, Rt = 20 m signal curves by about 20 dB in the figures. 

Thus, marginal but adequate radio comunication may still be possible 

at frequencies between about 300 kHz and 1 MHz when one of the units is 

located in an adjacent entry. 

Other analyses, by Arthur D. Little, Inc.,** also indicate that 

attenuation losses on trolley wirelrail transmission lines can increase 

by an order of magnitude if the line is heavily shunt loaded, as is the 

case for many trolley wirelrail lines. In such cases, loop-to-loopvia-the-line system longitudinal range will most likely be limited by 

the distance between discrete shunt loads such as power rectifiers. 

*Dushac, H.M., "TWO-Way Wireless Voice Communication System for Under 

ground Coal Mines1', Internal Report, Lee Engineering Division, 

Consolidation Coal Company, February, 1976. 

**Arthur D. Little, Inc. , "Improvements for Mine Carrier Phone Systems ,I' 

Report on Task 11, Task Order No. 2, Contract No. H0346045, May 1977. 

I 

Arthur D Little. Inc. 
In conclusion, to better characterize the behavior and determine 

the limits and the practicality of both loop-to-loop via-the-line and 

totally wireless loop-to-loop communications in mines, a measurement 

program has been defined and is being conducted in several coal mines 

for the Bureau of Mines, PMSRC, by a measurement and analysis team 

consisting of Spectra Associates/~ollins Radio Group on Contract 

H0366028 and Arthur D. Little, Inc., on Contract H0346045, Task Order 

No. 4. 

Arthur D Little Inc. 
(This page intentionally left blank) 

LI 

Arthur D Little Inc 
V, HYBRID NOISE CANCELLING DIVERSITY RECEIVING TECHNIQUE 

A. SUMMARY 

A hybrid noise cancelling diversity receiving technique has been 

examined that would try to capitalize on the high levels of both electric 

and magnetic field noise components present in many parts of a mine at 

the frequencies of interest below 1 MHz, Namely, if the instantaneous 

correlation is high enough between the noise E and H field components, 

it should be possible to improve the output signal-to-noise performance 

of a mine wireless system by cancelling a major portion of the H-field 

noise picked up in a loop antenna by properly combining it (manually or 

automatically) with the E-field noise picked up by a dipole or whip antenna that would be insensitive to both the H-f ield signals generated by 

the wireless transmitter and the ambient H-f ield noise, This selectivity 

between signal and noise would come about from the fact that the loop 

signal transmitters generate very little E field and are coupled to the 

receiver primarily through magnetic induction, Calculations to date indicate that potential signal-to-noise improvements between 10-15 dB may 

be possible when the E and H noise components are 80% to 90% correlated, 

a condition which may prevail in the vicinity of severe noise sources, 

Such a SIN increase could then result in extending the range of a wireless communications system or reducing the power to attain a given range, 

The next step required in the pursuit of this approach is a set of basic 

field measurements to determine whether the degree of correlation between 

the E and H noise field components is high enough to warrant a more rigorous analysis and further development effort. 

B . INTRODUCTION 

The review and analysis work treated in the previous chapters lead 

to the following comments and observations. No matter what equipment is 

used for mine wireless electromagnetic communications below about 1 MHz, 

one encounters three difficulties in underground mines. 

The first is that at the low and medium frequencies desired, the 

attenuation and field fall off in a homogeneous conductor-free region 

are very high because of the surrounding medium losses and the fact that 

Arthur D Little Inc. 
near zone conditions prevail for the transmitter and receiver. To date, 

because of power limitations in portable equipment, it has been difficult 

to provide communications for distances up to a few thousand feet even in 

ideal ground. Depending upon the investigator, the model he establishes 

and the assumptions made about system parameters, the signal strength has 

been calculated to vary somewhere between r~l'*e-~~ and r -3 e -ar over the 

ranges of interest, This is a formidable barrier to overcome no matter 

what the specific details of the system are. Even the favorable results 

obtained from mine wireless propagation tests in Consolidation Coal's 

Ireland mine indicate that the desired portable unit wireless range of 

1350 feet may only be attainable under the most favorable conditions of 

noise and medium conductivity. These results and a propagation model to 

fit them are presented in the Task F final report of this Contract* and 

have been sunnnarized in Chapter I1 of the present report. 

The second difficulty is that in addition to the weak signal one 

expects to receive at extreme range, there can be an extremely high background of electromagnetic noise coming in with the signal due to a variety 

of causes. This noise is generated by DC traction machines, AC to DC converters, a variety of tools, mining equipment, pumps, compressors, etc., 

all of which are working in an active mine, Measurement of the noise power 

at various frequencies indicate that it is of significant level up to well 

above most of the frequencies one would like to use to establish a minerto-miner voice-modulated EM communication system in the VLF to MF bands. 

The third difficulty is that in an operating mine both signals and 

noise will be carried to most places where miners are located, as a result 

of the random excitation of trolley wires, pipes, tracks, telephone pairs, 

and the like. 

Because of this chaotic physical environment, the receiving antenna 

of a wireless induction system receives a number of E and H components 

and reacts to a complicated vector summation of these, These components 

are derived from a number of sources, active and parasitic, and have a 

number of different properties. A few of these components may be listed: 

From nearby trolley wires there is an induction H field 

due to DC and AC power transients as well as trolley 

*Arthur D, Little, Inc., Task F--Final ~eport--"~ropagation of Radio Waves 

in Coal ~ines," Chapter IV, U,S. Bureau of Mines Contract H0346045, 

Task Order NO. 1, October 1975. 

124 - 

Arthur D Little. Inc 
phone currents around 100 kHz and its harmonics, 

From distant sources there may be plane waves of natural and man-made interference in which E and H are 

related by a characteristic impedance. 

From a nearby wireless transmitter there may be induction field components of correlated E and H arriving 

from an arbitrary direction with an arbitrary 

polarization, 

From trolley wires there may be strong E fields due 

to arcs and switching transients not necessarily 

associated with any H field. 

There can be induced voltages and currents from any 

of these primary sources inductively coupled to wires, 

cables and piping which will result in secondary coupling and a repeat and complication of all the factors 

we have just cited. 

Measurements made by the National Bureau of Standards* in several coal 

mines substantiate these generalities both with respect to noise power 

level and its variation with time and location, 

Finally, we observe that none of these conditions or any others similar to them are stable, Mining operations progress, wires are changed 

and removed, power systems are altered, pipes, pumps and trailing cables 

will be shifted, Last, but not least, the spatial relationship of the 

transmitter and receiver for a miner-to-miner communication system can 

vary according to the movement of the men and equipment as well as their 

particular physical positions, 

*Bensema, WoDo, Me Kanda, and Jew, Adams (1974a), Electromagnetic Noise 

in Robena NO, 4 Coal Mine, Nat, Bur, Stand. (U.S.), Tech. Note 654, 

194 pp,, April1974, SD Cat, NO, C13,46:654, Sup, of Doc., U.SO Government Printing Office, Washington, DOCo 20402 (U,S, Bureau of Mines Contract H0133005). 

Kanda, Me, Jewo Adams, and WoDo Bensema (1974c), Electromagnetic Noise 

in McElroy Mine, Nat, Bur, Stand, (U.S.), NBSIR 74-389, 170 pp,, June 

1974, order under SD Cat, NO, C13, Sup, of Doc,, U,S, Government 

Printing Office, Washington, DOCo 20402 (U,S, Bureau of Mines Contract 

H0133005). 

Bensema, WoD,, M, Kanda, and Jewo Adams (1974b), Electromagnetic Noise 

in Itmann Mine, Nat, Bur, Stand, (U.S.) , NBSIR 74-390, 112 pp., June 1974, 

order under SD Cat, No. C13, Sup, of DOC,, U,S, Government Printing 

Off ice, Washington, DOC, 20402 (U, S, Bureau of Mines Contract ~0133005). 

125 Arthur D Little Inc. 
Up to now methods recommended for combatting mine electromagnetic 

noise as it affects induction or wireless radio systems have mostly involved: modulation techniques, signal bandwidth reduction, voice processing or signal coding, optimizing carrier frequency selection, achieving 

maximum power efficiency. These are what one might call "whole signal" 

techniques and do not make use of the fact that before the total signal 

is converted to a single current in a receiver, there are noise components of that signal which may be separated from the information carrying portion. Specifically, by the use of separate antennas, sensitive 

independently to E and H fields, perhaps separately spaced in special 

configurations, and fed to phasing and balancing networks, a substantial 

increase in signal-to-noise ratio may be achievable even before the whole 

signal processing techniques are applied within the receiver, 

The rest of this chapter is devoted to the description and analysis 

of a signal reception technique that should be subjected to test and 

assessment to determine whether it may be feasible to realize an overall 

system performance improvement when in the presence of mine electromagnetic noise. The technique involves making use of the possible correlations which may exist between the electric and magnetic field vectors of 

the mine electromagnetic noise, If these correlations can be demonstrated, 

there is real possibility of constructing a receiver which, in its simplest form, receives and combines outputs from two separate antennas; 

one primarily sensitive to the E-field, and the other to the H-field as 

shown in Figure 5-1, This receiver would combine the antenna outputs of 

these separate sensors in such a way (vectorially and perhaps adaptively) 

that the noise picked up by the E-field antenna would be used to cancel 

the correlated noise component of the total output received by the loop 

or H-field antenna, thereby enhancing the signal-to-noise ratio of the 

final output, This should be possible if the signal-to-noise ratio at 

each antenna is different and the noise components are highly correlated. 

Thus, by amplitude, phase, and possibly directional adjustments, it may 

be possible to improve, before any further processing steps, the signalto-noise ratio at the receiver output terminals. This proposed technique, 

is in its simplest form, a diversity reception scheme using E- and H-field 

components which can be received separately and independently by selective 

E- and H-field sensors, 

Arthur D Little. lnc. 
Magnetic Field Electric Field 

- 

( SH+ NH+ iH) (3,. ME+&) 

0 Loop 

(Gain and Phase Control) 

A B 

that this is 

Zero 

Source: Arthur D. Little, Inc. 

FIGURE 5-1 HYBRID NOISE CANCELLING DIVERSITY RECEIVING 

TECHN IQUE (Simplified Basic Operation) 

Arthur D Little Inc. 
C , S IGNAL-TO-NOISE RATIO IMPROVEPENT ANALYSIS 

1, Nomenclature 

Suppose that an antenna which is only sensitive to the H field picks 

up a signal voltage combined with noise and suppose that an adja- H H 

cent E antenna picks up a signal combined with noise % (The bars E E' 

over the letters denote a vector quantity), If the two noise voltages 

are partly correlated with each other, we can let n = the uncorrelated H 

part of the H noise and n = the uncorrelated part of the E noise. Let E - 

the parts which are correlated be and NEC, respectively. Since they HC 

are correlated, it should be possible to cancel one with the other, One 

way to achieve this is to pass the output of the E antennas through a 

variable gain amplifier and a variable phase changer, This enables us 

to manipulate the magnitude and phase of the voltage received from the 

E-antenna prior to summation of the E and H inputs to the receiver as 

depicted in Figure 5-1, 

The presence of an electric field signal is not necessary for E 

this technique to be applied, Only the presence of noise is required on 

the E-field antenna, In fact, the following analysis implicitly assumes 

that if an E-signal is present, a means must be provided to independently 

sense and process the E-field noise without interference from the E-field 

signal, Otherwise, whenever the signal-to-noise ratio on the E-antenna 

was much greater than about 1, the noise canceller would try to cancel 

the correlated E and H signal outputs, thereby drivingmthe signal level 

to a minimum instead of the noise, The remainder of this chapter is concerned with the examination of the feasiability and potential utility of 

such a noise cancelling technique from an analytic viewpoint, and not on 

the conceiving of specific manual or automatic means of implementing it 

in practical hardware. The analysis also assumes a substantial degree 

of correlation between the magnetic and electric field noise components, 

This must also be verified by measurements, 

2. Signal-to-Noise Ratio 

We can write the voltage at the E antenna as: 

Arthur D Little Inc 
After passing through the amplifier and phase changer this voltage becomes: 

- 

VE = G(SE + NEc + iiE) (2) 

where : 

G = ge.ii!' (3) 

g = the adjustable gain (or attenuation) of the amplifier 

$ = the adjustable phase angle of the phase changer. 

The voltage from the H antenna can be written: 

If we sum the two voltages (2) and (4) above by passing them through a 

summing network we get the voltage: 

The correlated noise portion of the voltage is NHc + sEC). From Figure 2 

5-2 it is seen that IFHc + zECl can be written as 

where A = an arbitrary constant phase angle between the correlated 

portions of the E and H noise vectors 

and @=++A. (7) 

The convention lGHc I = NHC is used to denote vector magnitude. If we 

assume, as we are perfectly free to assume, that standard impedances 

apply throughout, (6) represents the correlated noise power, PN . Now, - 

assuming it possible to adjust g so that 

and adjust J, so that 

(8b) 

Arthur D Little Inc. 
-- 

GN E N~cCos @ 

Source: Arthur D. Little, Inc. 

FIGURE 5-2 CORRELATED NOISE VECTOR DIAGRAM 

n 

Arthur D Little lnc. 
expression (6) becomes zero. Namely, the correlated noise would be perfectly cancelled by making such that 

In practice, the presence of uncorrelated noise components would set a 

lower limit to the degree of cancellation of even the correlated components. 

The intent here is to assess whether the potential range improvement 

obtainable under favorable cancelling situations warrants a more rigorous 

experimental and analytical investigation. 

When the correlated noise is cancelled as defined in (9), the summed voltage (5) becomes: 

where now has the special value assigned by (8) and (9) above. Suppose 

that, under these circumstances, the net phase angle remaining between zH 

and zE is 8. Then as in (6) above we can write 

This is the signal power at the output of the summing network. Since n H 

and SE are uncorrelated, the noise portion of the voltage (1) above has 

a power P of 

n 

Therefore the signal-to-noise ratio R of (10) is given by C 

If the H antenna is used alone, the signal-to-noise power ratio reduces 

Arthur D Little. lnc. 
Dividing (13) by (14) and substituting (8a) for the value of g, gives a 

rational measure of the signal-to-noise ratio improvement obtained by this 

noise cancelling technique. Calling this improvement ratio the Improvement Factor, I, it can be shown that: 

where : 

REC = (SE/NEC), E signal to E correlated noise ratio 

RHC = (SHINHC), H signal to H correlated noise ratio 

a = (nH/NHC), H uncorrelated to correlated noise ratio 

0 = (nE/NEC). E uncorrelated to correlated noise ratio 

3. Circle Diagram Interpretations 

An insight into the behavior of the Improvement Factor I as a function of the relative signal and noise levels and the net phase angle 8 

between the E and H signal outputs can be obtained by using a circle diagram which occurs repeatedly in engineering, If we regard 8 as the angle 

between a unit vector i and the vector p = G/Qc, we draw the diagram RC 

of Figure 5-3 in which the length of the vector. OQ, is seen to be: 

As 0 varies, the locus of the point Q is a circle of radius pRC = IR~~/R~~(, 

centered on (1.0). Figure 5-4 illustrates how (0~)~ expressed in dB behaves as a function of e and the relative values of REC and RHC. Figure 

5-4 shows that if the E-field signal-to-correlated noise ratio REC is 

greater than twice the H-field signal-to-correlated noise ratio RHC, the 

Improvement Factor I will be increased by the presence of the E-signal 

(assuming that the gain g is set independently of sE), However, if REC 

is less than twice %, the presence of SE can cause a reduced value of I 

over that attainable in the absence of SE; namely, whenever the resultant 

phase angle 0 falls between 90' and 270'. 

Arthur D Little. Inc -. I 
Arthur D Little. Inc. 
Arthur D Little Inc. -1 
Now let another circle of radius p be drawn centered at the origin Nn 

as in Figure 5-5, 

The Improvement Factor, I, now shows directly on the figure as the square 

of the ratio of OQ to OM. 

Points P in Figure 5-5 have a special interest. As shown in Figure 5-6, 

these points give the values 8 for which I is unity; i,e,, no improvement. 0 

If we consider 8 as an equally distributed random variable, then Figure 

5-6 shows that the probability p that I - > 1, is given by, p = 1 eel In. Further examination of Figure 5-5 reveals that when an E-signal S is not E - 

present, p = 0 and OQ reduces to unity, so that the Improvement Factor RC 

depends only on the degree of noise correlation; namely 

When the degree of correlation between the E and H components is 

very small; i. e. , when NH, NE -t 0, the gain setting g defined by (8a) will 

be indeterminate. In that case the required gain and phase adjustment 

reduces mathematically to unity gain with zero phase shift. Thus (20) 

would reduce to 

which reveals a decrease in net signal-to-noise ratio compared to reception on only the loop antenna, Conversely, when the degree of correlation between the E an£ H field noise components is large; i.e., a, 6 small, 

(20) becomes 

Arthur D Little Inc. 
Source: Arthur D. Little, Inc. 

FIGURE 5-5 CIRCLE DIAGRAM FOR SIGNAL AND NOISE FACTORS OF I 

Source: Arthur D. Little, Inc. 

FIGURE 5-6 CIRCLE DIAGRAM INDICATING REGION WHERE I < 1 

136 -. 

Arthur D Little Inc 
Examination of (22) reveals the following. In the ideal case of 

completely correlated noise, a, 6 = 0, the noise cancellation would be 

perfect, giving infinite improvement. If the H-antenna output contains 

only correlated noise, then a = 0, and (22) reduces to 

which gives significant improvement whenever is small. If the E-antenna 

output contains only correlated noise, then 6 = 0 and (22) reduces to 

leading to significant improvement whenever a is small. Finally, if the 

ratio of uncorrelated-to-correlated noise is non-zero but of similar value 

for both E and H antenna outputs; i.e., a = %INH 2 nE/NE = 6, (22) reduces 

approximately to 

giving half the improvement of the B = 0 condition. 

4. Expected Values 

It should be emphasized that the circle diagrams of Figure 5-3 and 

5-5 and the plot of Figure 5-4 have been included here to illustrate the 

role played by the variable 8. Since we lack experimental data, we will 

deal primarily with the expected value of 8. In view of the wide variety 

of coupling structures and noise sources likely to be encountered in mines, 

the only rational assumption we can make in regard to 8 is that it is an 

equally distributed random variable. Consequently, the expected value of 

cos 8 in (15) is 0, the expected value of 8 can be taken as ~/2, and the 

expected value of I is given by 

Arthur D Little Inc 
This is the statistical measure of performance averaged over all situations encountered, Expressing (26) in dB we get 

<I> (dB) = 10 log,(, <I> 

Figure 5-7 presents plots of the "average" improvement factor <I> in dB, 

as a function of the H-field correlated-to-uncorrelated noise factor, lla, 

for two representative values of the relative correlation ratio Bla, and 

five values of the ratio of signal-to-noise ratios on the E and H antennas, 

The figure illustrates that, in the absence of an E-f ield signal, when the 

E and H field noises enjoy the same degree of correlation (B/a = I), significant improvement occurs as the correlated-to-uncorrelated noise ratio 

l/a increases; i.e., <I> = 17 dB for a = 0.1, l/a = 10. Only a marginal 

3 dB additional improvement is obtained if the E-field noise is much more 

correlated than the H-field noise (B/a = 0.1). It is also seen that the 

presence of an E-field signal (if it is not allowed to interfere with 

the noise cancelling process) may also on the average result in a net 

improvement, depending on the value of the relative signal-to-noise ratio 

factor (REC/%C)* 

The plots of <I> in Figure 5-7 assume an average value of IT/^ for 8. 

If 8 has some other value, the expression for <I> given by equations (27) 

would need a correction term of Q dB added to it, where Q is given by 

I 1 + (REC/RHC) + 2(REC/RHC) case 

Q = 10 log,, 1. 

to represent the improvement I in dB for a specific angle 8, Figure 5-8 

plots Q against 8 for various values of R EC/%C , This figure gives the 

number of dB which must be added to the values of Figure 5-7 to find I 

for a particular value of 8, and illustrates that the presence of an Efield signal can be deterimental when the resultant phase angle 8 obtained 

from the noise cancelling procedure falls between 90' and 270°. 

138 

Arthur D Little Inc 
For Pla = 

0.1 1 .o 

Source: Arthur D. Little, Inc. 

FIGURE 5-7 EXPECTED VALUE OF THE IMPROVEMENT FACTOR < I > 

IN dB VERSUS l/a AND (REC/RHC) FOR TWO VALUES OF PIa 

(0.1, 1 .O) AND <8> = n/2 

Arthur D Little. lnc 
FIGURE 5-8 CORRECTION TERM Q TO <I> VERSUS PHASE ANGLE 

BAND RATIO REC/RHC 
5. Boundary Plots and Performance 

Another instructive way to plot the dependence of <I> on the degree of 

noise correlation and the signal-to-correlated noise ratios at the E and H 

antennas is in the form of boundary plots. Namely, choose desired Improvement Factor values <I> and plot the parameter values need to achieve these 

Improvement Factors. As an example, Figure 5-9 presents the combination of 

values of (REC/k) and l/a to achieve four different degrees of s/N improvement <I> under noise conditions where f3/a = 1. Note the key role played by 

l/a, which is a measure of the degree of noise correlation. In the absence 

of an E-signal, if l/a is larger than about 6 or 7, improvements in excess of 

12 dB can be achieved; improvements that could mean the difference between 

marginal or no communications and satisfactory communication in a high ambient electrical noise environment. To substantially reduce this requirement 

on the value of l/a, (REC/%c) must exceed unity, a condition that presently 

appears improbable for inductively coupled wireless communication systems 

in a mine environment. The above analysis indicates that the degree of 

noise correlation in mine environments is of critical importance, and therefore needs to be established for the frequency band of interest. 

For the sake of simplicity in the above analysis, we have used a simple 

phase shif ter and variable gain stage to illustrate the noise cancellation 

process. The noise measurements should examine not only the variables of 

gain and phase to accomplish cancellation, but also the spacing and relative orientation between antennas, since it is anticipated that these geoib~etrical factors may also play an important role in achieving discrimination 

against the adverse effects of mine noise. 

It is not yet clear exactly how such an adaptive noise cancelling 

process to maximize output signal-to-noise ratio will be implemented. 

Completely automatic methods to quite rudimentary manual methods that can 

be switched in and out at will, can bc conceived. If the method is simple 

but incapable of distinguishing between signals and noise, the presence of 

an appreciable E-signal will clearly be a liability. However, when a 

strong E-signal is present, adequate system performance should be obtainable 

simply by switching the canceller out of the receive circuits, and choosing 

Arthur D Little. Inc. 
- For P/a = 1 

- 

- 

- 

S/N l mprovement 

Factor <I> 

Source: Arthur D. Little, Inc. 

FIGURE 5-9 BOUNDARY PLOT OF RE~/RHC VERSUS Ila 

FOR SPECIFIED IMPROVEMENT FACTORS 

<I> WITH P/a = 1 

LArthur D Little Jnc 
the channel with the best perceived signal-to-noise ratio, An automatic 

system capable of distinguishing between signals and noise, and then 

operating only on the noise, may prove to be too complex for the intended 

portable application for use in mines unless the noise is found to exhibit 

some highly favorable distinguishing features that can be easily utilized, 

The conditions which appear most favorable for using a simple means 

to achieve significant improvements through noise cancelling are those in 

which both E and H field noise components are present and highly correlated with each other, and in which the E-field signal-to-correlated noise 

is negligible compared to the H-field signal-to-correlated noise ratio. 

Without measurements of the degree of correlation between the E and H 

field components of mine electromagnetic noise, further analysis is probably a waste of time, The next section outlines a modest series of field 

measurements for quickly estimating the degree of correlation between 

in-mine E- and H-field noise, 

Do RECOMMENDED FIELD MEASUREMENTS 

To see if some form of noise cancelling diversity reception involving 

instantaneous correlation between E and H fields as well as direction of 

arrival is possible, we recommend that a modest number of basic measurements be made in an environment which simulates to a first order the 

type of industrial RFI environment which one encounters in operating 

mines. The first and simplest is to obtain a measure of the normalized 

cross-correlation between the E and H vectors of the noise field. 

If this correlation is found to be high, then in a magnetic induction wireless link, we can hope to cancel the H-noise picked up on a 

loop antenna by using the E-noise picked up on a dipole or whip antenna 

which would be insensitive to the H-field generated by the signal transmitting device, This selectivity comes about from the fact that the 

transmitter will utilize a loop antenna which generates very little E 

field, and which is "connected" to the receiver through magnetic induction. 

A further extension of the use of separate E and H sensitive antennas 

is the testing of the potential benefits of signal and noise differentiation based on sensitivity to antenna orientation. 

Arthur D Little Inc. 
To assess the feasibility of the above discussed noise cancelling 

technique we recommend the following set of field measurements: 

From existing laboratory components and equipment, 

assemble an experimental test receiving system 

operating in the 15 to 300 kHz range which will 

simultaneously display on an oscilloscope face a 

measure of the instantaneous cross-correlation 

of the normalized voltages received by independent 

E-field and H-field sensitive antennas. This portable equipment should be checked and calibrated 

in the laboratory and simple procedures for operation and adjustment established. 

Make arrangements to allow access to a nonaine 

industrial site which uses heavy electrical equipment, in order to measure the E- and H-fields produced by traction motors, welders, etc, A trolley 

car barn or subway switch yard typifies such an 

environment and should provide an adequate representation of a mine environment. If desired, these 

measurements could be followed by in-mine measurements to confirm the findings for the above industrial site. 

Take themeasurements at thesitewithdifferent 

equipment sensitivities and antenna orientations 

in various locations to determine some empirical 

rules as to the degree and constancy of the correlation between E- and H-field noise components in 

real time, 

If a strong correlation effect is discovered, perform additional measurements with the same test 

equipment and a signal source to demonstrate that 

a signal-to-noise improvement is possible. This 

critical experiment should involve the temporary 

establishment at some distance from the measuring 

144 CI* 

Arthur D Little Inc 
equipment of a simple CW transmitting loop to provide 

the test signal, 

Upon the completion of these tests, prepare a test 

report describing the results of this work, including 

a series of conclusions and recommendations relating 

to the next steps to be taken, 

The purpose of this modest program is to perform in the simplest 

and most rapid manner a series of basic measurements to explain the 

physics of the situation. If these tests and measurements prove successful, a follow-on effort aimed at prototype equipment design and development can be pursued, 

Arthur D Little fnc. 
ATHENS 

BRUSSELS 

LONDON 

PARIS 

RIO DE JANEIRO 

TORONTO 

WIESBADEN