# Patent

## Title
Dual link power converter

## Abstract

Methods and systems for transforming electric power between two or more portals using multiple power modules. Any or all portals can be DC, single phase AC, or multi-phase AC. Individual power modules comprise a plurality of bi-directional conducting and blocking semiconductor switches, and an inductor and parallel capacitor (reactance). The switches alternately connect the reactance between said portals, such that energy is transferred into the inductor from one or more input portals and/or phases, then transferred out of the inductor to one or more output portals and/or phases, with said parallel capacitor facilitating “soft” turn-off, and with any excess inductor energy being returned to the input. Dual power modules can operate 90 degrees out of phase. This configuration allows use of the same I/O filter capacitors as with a single power module, while achieving twice the total power produced by the power converter, reducing ripple voltage and doubling ripple frequency.

## Background
CROSS-REFERENCE
Priority is claimed from U.S. Provisional App. No. 61/765,110, filed Feb. 15, 2013, which is hereby incorporated by reference.
Priority is claimed from U.S. patent application Ser. No. 13/859,265, filed Apr. 9, 2013; which is a continuation of U.S. patent application Ser. No. 13/214,575, filed Aug. 22, 2011; which is a continuation of U.S. patent application Ser. No. 12/479,207, filed Jun. 5, 2009, and issued as U.S. Pat. No. 8,300,426 on Oct. 30, 2012; which is a continuation of U.S. patent application Ser. No. 11/759,006, filed Jun. 6, 2007, and issued as U.S. Pat. No. 7,599,196 on Oct. 6, 2009; which claims priority from U.S. Provisional App. No. 60/811,191, filed Jun. 6, 2006; all of which are hereby incorporated by reference.


BACKGROUND
The present application relates to electric power conversion, and more particularly to multi-port power converters.
Note that the points discussed below may reflect the hindsight gained from the disclosed inventions, and are not necessarily admitted to be prior art.
A new kind of power converter was disclosed in U.S. Pat. No. 7,599,196 entitled “Universal power conversion methods,” which is incorporated by reference into the present application in its entirety. This patent describes a bidirectional (or multidirectional) power converter which pumps power into and out of a link inductor which is shunted by a capacitor.
The switch arrays at the ports are operated to achieve zero-voltage switching by totally isolating the link inductor+capacitor combination at times when its voltage is desired to be changed. (When the inductor+capacitor combination is isolated at such times, the inductor's current will change the voltage of the capacitor, as in a resonant circuit. This can even change the sign of the voltage, without loss of energy.) This architecture has subsequently been referred to as a “current-modulating” or “Power Packet Switching” architecture. Bidirectional power switches are used to provide a full bipolar (reversible) connection from each of multiple lines, at each port, to the rails connected to the link inductor and its capacitor. The basic operation of this architecture is shown, in the context of the three-phase to three-phase example of patent FIG. 1, in the sequence of drawings from patent FIG. 12a to patent FIG. 12j. 
The ports of this converter can be AC or DC, and will normally be bidirectional (at least for AC ports). Individual lines of each port are each connected to a “phase leg,” i.e. a pair of switches which permit that line to be connected to either of two “rails” (i.e. the two conductors which are connected to the two ends of the link inductor). It is important to note that these switches are bidirectional, so that there are four current flows possible in each phase leg: the line can source current to either rail, or can sink current from either rail.
Many different improvements and variations are shown in the basic patent. For example, variable-frequency drive is shown (for controlling a three-phase motor from a three-phase power line), DC and single-phase ports are shown (patent FIG. 21), as well as three- and four-port systems, applications to photovoltaic systems (patent FIG. 23), applications to Hybrid Electric vehicles (patent FIG. 24), applications to power conditioning (patent FIG. 29), half-bridge configurations (patent FIGS. 25 and 26), systems where a transformer is included (to segment the rails, and allow different operating voltages at different ports) (patent FIG. 22), and power combining (patent FIG. 28).
Improvements and modifications of this basic architecture have also been disclosed in U.S. Pat. Nos. 8,391,033, 8,295,069, 8,531,858, and 8,461,718, all of which are hereby incorporated by reference.
The term “converter” has sometimes been used to refer specifically to DC-to-DC converters, as distinct from DC-AC “inverters” and/or AC-AC frequency-changing “cycloconverters.” However, in the present application the word converter is used more generally, to refer to all of these types and more, and especially to converters using a current-modulating or power-packet-switching architecture.
Current-modulating power converters can convert DC to DC, DC to AC, and AC-AC, and are suitable for applications including, e.g., line power conditioners, battery chargers, hybrid vehicle power systems, solar power systems, motor drives, and utility power conversion.
Numerous techniques have been proposed for electronic conversion of electric power from one form into another. A technique in common commercial usage for operating three phase induction motors at variable frequency and voltage off of fixed frequency and voltage utility power is the AC-DC-AC technique of the input diode bridge, DC-link capacitor, and the output active switch bridge, under pulse-width modulation control. However, such an AC-DC-AC conversion technique is problematic, as explained in U.S. Pat. No. 7,599,196.
SUMMARY
The inventor has discovered that various advantageous synergies can be achieved by using multiple soft-switched power modules connected in parallel. In some preferred embodiments (but not necessarily in all), individual power modules comprise a plurality of bi-directional conducting and blocking semiconductor switches, and an inductor and parallel capacitor (reactance). The switches alternately connect the reactance between said portals, such that energy is transferred into the inductor from one or more input portals and/or phases, then transferred out of the inductor to one or more output portals and/or phases, with said parallel capacitor facilitating “soft” turn-off, and with any excess inductor energy being returned to the input.
The disclosed innovations, in various embodiments, provide one or more of at least the following advantages. However, not all of these advantages result from every one of the innovations disclosed, and this list of advantages does not limit the various claimed inventions.

    
    
        Reduces ripple voltage;
        increases ripple frequency;
        increases power produced over a given amount of I/O filter capacitors;
        provides more uniform current draw;
        provides more uniform current supply;
        reduces per unit filtering requirement for a power converter;
        high-bandwidth active control ability—more so than resonant or voltage-source or current-source converters;
        design versatility;
        power efficiency;
        optimal use of device voltage ratings;
        high power density converters;
        high power quality (low input and output harmonics with minimal filtering);
        voltage buck and boost capability;
        bi-directional, or multi-directional power transfer capability;
        high frequency power transformer capability, allowing for compact active transformer and full galvanic isolation if desired;
        input-output isolation even without a transformer, allowing for output with no common-mode voltage; and
        moderate parts count resulting from absence of auxiliary power circuits for snubbing.
    
    





BRIEF DESCRIPTION OF THE DRAWINGS
The disclosed inventions will be described with reference to the accompanying drawings, which show important sample embodiments and which are incorporated in the specification hereof by reference, wherein:
FIG. 1 schematically shows an example of a power converter with dual, parallel power modules.
FIG. 2 shows an example of a current waveform from dual parallel power modules.
FIG. 3A shows a simplified schematic of a sample power converter.
FIG. 3B shows sample voltage and current waveforms for a power cycle of a sample power converter.
FIG. 3C shows an exemplary finite state machine for one sample control architecture.
FIGS. 3D, 3E, and 3F show sample embodiments of output and input voltages.
FIG. 3G shows one sample embodiment of a bidirectional switch.
FIG. 3H shows one sample embodiment of a bidirectional current-modulating power converter.
FIGS. 3I, 3J, 3K, 3L, 3M, 3N, 3O, 3P, 3Q, and 3R show sample voltage and current waveforms on an inductor during a typical cycle while transferring power at full load from input to output.
FIG. 3S shows voltage and current waveforms corresponding to the full power condition of FIGS. 3I-3R, with the conduction mode numbers corresponding to the mode numbers of FIGS. 3I-3R.
FIG. 3T shows an embodiment of the present inventions with a full bridge three phase cycle topology, with controls and I/O filtering, including a three phase input line reactor as needed to isolate the small but high frequency voltage ripple on the input filter capacitors from the utility.
FIG. 3U shows an embodiment of the present inventions with DC or Single Phase portals.
FIG. 3V shows an embodiment of the present inventions with a transformer/inductor.
FIG. 3W shows an embodiment of the present inventions in a four portal application mixing single phase AC and multiple DC portals, as can be used to advantage in a solar power application.
FIG. 3X shows an embodiment of the present inventions in a three portal application mixing three phase AC portals and a DC portal, as can be used to advantage in a Hybrid Electric Vehicle application.
FIG. 3Y shows an embodiment of the present inventions as a Half-Bridge Buck-Boost Converter in a Single Phase AC or DC Topology with BCBS.
FIG. 3Z show a sample embodiment in a Half-Bridge Buck-Boost Converter in a Three Phase AC Topology with BCBS.
FIG. 3AA shows a sample embodiment in a single phase to three phase synchronous motor drive.
FIG. 3BB shows a sample embodiment with dual, parallel, “power modules”, each of which consists of 12 bi-directional switches and a parallel inductor/capacitor. More than two power modules can of course be used for additional options in multiway conversion.
FIG. 3CC shows an embodiment of the present inventions as a three phase Power Line Conditioner, in which role it can act as an Active Filter and/or supply or absorb reactive power to control the power factor on the utility lines.
FIG. 3DD shows a sample schematic of a microgrid embodiment.
FIG. 3EE shows another sample embodiment of a microgrid.
FIG. 4A shows an example of a current waveform from an output capacitor using a single power module.
FIG. 4B shows an example of a current waveform from an output capacitor using dual parallel power modules.
FIG. 5 shows an example of a current waveform from an output capacitor using dual parallel power modules.
FIG. 6 schematically shows an example of a Full-Bridge Buck-Boost Converter in a Three Phase AC Full Cycle Topology with Bi-directional Conducting and Blocking Switches (BCBS).



DETAILED DESCRIPTION OF SAMPLE EMBODIMENTS
The numerous innovative teachings of the present application will be described with particular reference to presently preferred embodiments (by way of example, and not of limitation). The present application describes several inventions, and none of the statements below should be taken as limiting the claims generally.
DEFINITIONS
Anchoring—Using a switch to fix the voltage of one end of the link of a line voltage. Any change in link voltage will occur on the other end of the link.
Direct Anchoring—Leaving one switch of line pair closed after a charge transfer is complete to anchor the voltage of one end of the link to the line voltage.
Indirect Anchoring—Anchoring that occurs at the start of a charge transfer one the change in link voltage cause one switch to conduct and anchor that end of the link to the line voltage.
Dominant Phase—The phase of the three phase port that has the largest amount of charge to be transfer to the link.
FPGA—Field programmable gate array.
GFDI—Ground fault detection and interruption.
Islanding—When part of a power system consisting of one or more power sources and loads that is, for some period of time, is separated from the rest of the system.
Link—Inductor and capacitor pair that transfer energy between input and output line pairs.
Line pair—Two lines of a port that can transfer energy to or from the link.
Line pair switches—The bidirectional switches that connect a line pair to the link. The switches are composed of two IGBT in series with parallel diodes.
Microgrid—A small power grid to deliver power from a converter to local loads. The converter is the only power source of the microgrid.
MPPT—Maximum Power Point Tracking, algorithm to maximize the amount of power from a photovoltaic array
Referring initially to FIG. 3H, illustrated is a schematic of a sample three phase converter 100 that illustrates the operation of a power-packet-switching converter. The converter 100 is connected to a first and second power ports 122 and 123 each of which can source or sink power, and each with a line for each phase of the port. Converter 100 can transfer electric power between said ports while accommodating a wide range of voltages, current levels, power factors, and frequencies between the ports.
The first port can be for example, a 460 VAC three phase utility connection, while said second port can be a three phase induction motor which is to be operated at variable frequency and voltage so as to achieve variable speed operation of said motor. The present inventions can also accommodate additional ports on the same inductor, as can be desired to accommodate power transfer to and from other power sources and/or sinks, as shown in FIGS. 3W and 3X.
Referring to FIG. 3H, converter 100 is comprised of a first set of electronic switches S1u, S2u, S3u, S4u, S5u, and S6u that are connected between a first line 113 of a link inductor 120 and each phase, 124 through 129, of the input port, and a second set of electronic switches S1l, S2l, S3l, S4l, S5l, and S6l that are similarly connected between a second line 114 of link inductor 120 and each phase of the output port. A link capacitor 121 is connected in parallel with the link inductor, forming the link reactance. Each of these switches is capable of conducting current and blocking current in both directions, as seen in e.g. FIG. 3G. Many other such bi-directional switch combinations are also possible.
The converter 100 also has input and output capacitor filters 130 and 131, respectively, which smooth the current pulses produced by switching current into and out of inductor 120. Optionally, a line reactor 132 can be added to the input to isolate the voltage ripple on input capacitor filter 131 from the utility and other equipment that can be attached to the utility lines. Similarly, another line reactor, not shown, can be used on the output if required by the application.
For illustration purposes, assume that power is to be transferred in a full cycle of the inductor/capacitor from the first to the second port, as is illustrated in FIG. 3S. Also assume that, at the instant the power cycle begins, phases Ai and Bi have the highest line to line voltage of the first (input) port, link inductor 120 has no current, and link capacitor 121 is charged to the same voltage as exists between phase Ai and Bi. The controller FPGA 1500, shown in FIG. 3T, now turns on switches S1u and S2l, whereupon current begins to flow from phases Ai and Bi into link inductor 120, shown as Mode 1 of FIG. 3I.
FIG. 3S shows the inductor current and voltage during the power cycle of FIGS. 3I-3R, with the Conduction Mode sequence 1300 corresponding to the Conduction Modes of FIGS. 3I-3R. The voltage on the link reactance remains almost constant during each mode interval, varying only by the small amount the phase voltage changes during that interval. After an appropriate current level has been reached, as determined by controller 1500 to achieve the desired level of power transfer and current distribution among the input phases, switch S2l is turned off.
Current now circulates, as shown in FIG. 3J, between link inductor 120 and link capacitor 121, which is included in the circuit to slow the rate of voltage change, which in turn greatly reduces the energy dissipated in each switch as it turns off. In very high frequency embodiments of the present inventions, the capacitor 121 can consist solely of the parasitic capacitance of the inductor and/or other circuit elements. (Note that a similar process is shown in FIG. 3O.)
To continue with the cycle, as shown as Mode 2 in FIG. 3K and FIG. 3S, switch S3l is next enabled, along with the previously enabled switch S1u. As soon as the link reactance voltage drops to just less than the voltage across phases Ai and Ci, which are assumed for this example to be at a lower line-to-line voltage than phases Ai and Bi, switches S1u and S3l become forward biased and start to further increase the current flow into the link inductor, and the current into link capacitor temporarily stops.
The two “on” switches, S1u and S3l, are turned off when the desired peak link inductor current is reached, said peak link inductor current determining the maximum energy per cycle that can be transferred to the output. The link inductor and link capacitor then again exchange current, as shown if FIG. 3J, with the result that the voltage on the link reactance changes sign, as shown in graph 1301, between modes 2 and 3 of FIG. 3S. Now as shown in FIG. 3L, output switches S5u and S6l are enabled, and start conducting inductor current into the motor phases Ao and Bo, which are assumed in this example to have the lowest line-to-line voltages at the present instance on the motor.
After a portion of the inductor's energy has been transferred to the load, as determined by the controller, switch S5u is turned off, and S4u is enabled, causing current to flow again into the link capacitor. This increases the link inductor voltage until it becomes slightly greater than the line-to-line voltage of phases Ao and Co, which are assumed in this example to have the highest line-to-line voltages on the motor. As shown in FIG. 3M, most of the remaining link inductor energy is then transferred to this phase pair (into the motor), bringing the link inductor current down to a low level.
Switches S4u and S6l are then turned off, causing the link inductor current again to be shunted into the link capacitor, raising the link reactance voltage to the slightly higher input line-to-line voltage on phases Ai and Bi. Any excess link inductor energy is returned to the input. The link inductor current then reverses, and the above described link reactance current/voltage half-cycle repeats, but with switches that are complimentary to the first half-cycle, as is shown in FIGS. 3N-3R, and in Conduction Mode sequence 1300, and graphs 1301 and 1302. FIG. 3O shows the link reactance current exchange during the inductor's negative current half-cycle, between conduction modes.
Note that TWO power cycles occur during each link reactance cycle: with reference to FIGS. 3I-3R, power is pumped IN during modes 1 and 2, extracted OUT during modes 3 and 4, IN again during modes 5 and 6 (corresponding to e.g. FIG. 3P), and OUT again during modes 7 (as in e.g. FIG. 3Q) and 8. The use of multi-leg drive produces eight modes rather than four, but even if polyphase input and/or output is not used, the presence of TWO successive in and out cycles during one cycle of the inductor current is notable.
As shown in FIGS. 3I-3S, Conduction Mode sequence 1300, and in graphs 1301 and 1302, the link reactance continues to alternate between being connected to appropriate phase pairs and not connected at all, with current and power transfer occurring while connected, and voltage ramping between phases while disconnected (as occurs between the closely spaced dashed vertical lines of which 1303 in FIG. 3S is one example.
In general, when the controller 1500 deems it necessary, each switch is enabled, as is known in the art, by raising the voltage of the gate 204 on switch 200 above the corresponding terminal 205, as an example. Furthermore, each switch is enabled (in a preferred two gate version of the switch) while the portion of the switch that is being enabled is zero or reverse biased, such that the switch does not start conduction until the changing link reactance voltage causes the switch to become forward biased. Single gate AC switches can be used, as with a one-way switch embedded in a four diode bridge rectifier, but achieving zero-voltage turn on is difficult, and conduction losses are higher.
In FIG. 3T, current through the inductor is sensed by sensor 1510, and the FPGA 1500 integrates current flows to determine the current flowing in each phase (line) of the input and output ports. Phase voltage sensing circuits 1511 and 1512 allow the FPGA 1500 to control which switches to enable next, and when.
FIGS. 3I-3R shows current being drawn and delivered to both pairs of input and output phases, resulting in 4 modes for each direction of link inductor current during a power cycle, for a total of 8 conduction modes since there are two power cycles per link reactance cycle in the preferred embodiment. This distinction is not dependent on the topology, as a three phase converter can be operated in either 2 modes or 4 conduction modes per power cycle, but the preferred method of operation is with 4 conduction modes per power cycle, as that minimizes input and output harmonics.
For single phase AC or DC, it is preferred to have only two conduction modes per power cycle, or four modes per link reactance cycle, as there is only one input and output pair in that case. For mixed situations, as in the embodiment of FIG. 3X which converts between DC or single phase AC and three phase AC, there can be 1 conduction mode for the DC interface, and 2 for the three phase AC, for 3 conduction modes per power cycle, or 6 modes per link reactance cycle. In any case, however, the two conduction modes per power half-cycle for three phase operation together give a similar power transfer effect as the singe conduction mode for single phase AC or DC.
Another sample embodiment of the present inventions is shown in FIG. 3U, which shows a single phase AC or DC to single phase AC or DC converter. Either or both input and output can be AC or DC, with no restrictions on the relative voltages. If a port is DC and can only have power flow either into or out of said port, the switches applied to said port can be uni-directional. An example of this is shown with the photovoltaic array of FIG. 3W, which can only source power.
FIG. 3V shows a sample implementation of a Flyback Converter. The circuit of FIG. 3U has been modified, in that the link inductor is replaced with a transformer 2200 that has a magnetizing inductance that functions as the link inductor. Any embodiment of the present inventions can use such a transformer, which can be useful to provide full electrical isolation between ports, and/or to provide voltage and current translation between ports, as is advantageous, for example, when a first port is a low voltage DC battery bank, and a second port is 120 volts AC, or when the converter is used as an active transformer.
In the embodiments of the present inventions shown in FIGS. 3W and 3X, the number of ports attached to the link reactance is more than two, simply by using more switches to connect in additional ports to the inductor. As applied in the solar power system of FIG. 3W, this allows a single converter to direct power flow as needed between the ports, regardless of their polarity or magnitude.
Thus, in one sample embodiment, the solar photovoltaic array can be at full power, e.g. 400 volts output, and delivering 50% of its power to the battery bank at e.g. 320 volts, and 50% to the house AC at e.g. 230 VAC. Prior art requires at least two converters to handle this situation, such as a DC-DC converter to transfer power from the solar PV array to the batteries, and a separate DC-AC converter (inverter) to transfer power from the battery bank to the house, with consequential higher cost and electrical losses. The switches shown attached to the photovoltaic power source need be only one-way since the source is DC and power can only flow out of the source, not in and out as with the battery.
In the sample power converter of FIG. 3X, as can be used for a hybrid electric vehicle, a first port is the vehicle's battery bank, a second port is a variable voltage, variable speed generator run by the vehicle's engine, and a third port is a motor for driving the wheels of the vehicle. A fourth port, not shown, can be external single phase 230 VAC to charge the battery. Using this single converter, power can be exchanged in any direction among the various ports. For example, the motor/generator can be at full output power, with 50% of its power going to the battery, and 50% going to the wheel motor. Then the driver can depress the accelerator, at which time all of the generator power can be instantly applied to the wheel motor. Conversely, if the vehicle is braking, the full wheel motor power can be injected into the battery bank, with all of these modes using a single converter.
FIGS. 3Y and 3Z show half-bridge converter embodiments of the present inventions for single phase/DC and three phase AC applications, respectively. The half-bridge embodiment requires only 50% as many switches, but results in 50% of the power transfer capability, and gives a ripple current in the input and output filters which is about double that of the full bridge implementation for a given power level.
FIG. 3AA shows a sample embodiment as a single phase to three phase synchronous motor drive, as can be used for driving a household air-conditioner compressor at variable speed, with unity power factor and low harmonics input. Delivered power is pulsating at twice the input power frequency.
FIG. 3BB shows a sample embodiment with dual, parallel power modules, with each module constructed as per the converter of FIG. 3H, excluding the I/O filtering. This arrangement can be advantageously used whenever the converter drive requirements exceed that obtainable from a singe power module and/or when redundancy is desired for reliability reasons and/or to reduce I/O filter size, so as to reduce costs, losses, and to increase available bandwidth.
The power modules are best operated in a manner similar to multi-phase DC power supplies such that the link reactance frequencies are identical and the current pulses drawn and supplied to the input/output filters from each module are uniformly spaced in time. This provides for a more uniform current draw and supply, which can greatly reduce the per unit filtering requirement for the converter. For example, going from one to two power modules, operated with a phase difference of 90 degrees referenced to each of the modules inductor/capacitor, produces a similar RMS current in the I/O filter capacitors, while doubling the ripple frequency on those capacitors. This allows the same I/O filter capacitors to be used, but for twice the total power, so the per unit I/O filter capacitance is reduced by a factor of 2. More importantly, since the ripple voltage is reduced by a factor of 2, and the frequency doubled, the input line reactance requirement is reduced by 4, allowing the total line reactor mass to drop by 2, thereby reducing per unit line reactance requirement by a factor of 4.
FIG. 3CC shows a sample embodiment as a three phase Power Line Conditioner, in which role it can act as an Active Filter and/or supply or absorb reactive power to control the power factor on the utility lines. If a battery, with series inductor to smooth current flow, is placed in parallel with the output capacitor 2901, the converter can then operate as an Uninterruptible Power Supply (UPS).
FIG. 3A shows an example of a circuit implementing this architecture. In this example, one port is used for connection to the AC grid (or other three-phase power connection). The other is connected to a motor, to provide a variable-frequency drive.
In FIG. 3A, an LC link reactance is connected to two DC ports having two lines each, and to a three-phase AC port. Each line connects to a pair of bidirectional switches, such that one bidirectional switch connects the respective line to a rail at one side of the link reactance and the other bidirectional switch connects the line to a rail at the other side of the link reactance.
In one sample embodiment, voltage and current across a link reactance can be seen in, e.g., FIG. 3B. Link voltage waveform 1301 and link current waveform 1302 correspond to an arbitrary set of inputs and outputs. After a conduction interval begins and the relevant switches are activated, voltage 1301 on the link reactance remains almost constant during each mode interval, e.g. during each of modes 1-8. After an appropriate current level has been reached for the present conduction mode, as determined by the controller, the appropriate switches are turned off. This can correspond to, e.g., conduction gap 1303. The appropriate current level can be, e.g., one that can achieve the desired level of power transfer and current distribution among the input phases.
Current can now circulate between the link inductor and the link capacitor, which is included in the circuit to slow the rate of voltage change. This in turn greatly reduces the energy dissipated in each switch as it turns off. After the link voltage reaches appropriate levels for the next set of lines, the appropriate switches are enabled, and energy transfer between the port and the link continues with the next line pair.
A power converter according to some embodiments of this architecture can be controlled by, e.g., a Modbus serial interface, which can read and write to a set of registers in a field programmable gate array (FPGA). These registers can define, e.g., whether a port is presently an input, an output, or disabled. Power levels and operation modes can also be determined by these registers.
In some embodiments, a DC port preferably has one line pair, where each line pair is e.g. a pair of lines that can transfer energy to or from the link reactance through semiconductor switches. A three-phase AC port will always have three lines, and will often have a fourth (neutral), but only two are preferably used for any given power cycle (of the inductor).
Given three lines, there are three possible two-line combinations. For example, given lines A, B, and C, the line pairs will be A-B, B-C, and A-C.
Register values for each port can be used to determine the amount of charge, and then the amount of energy, to be transferred to or from each port during each conduction period. An interface then controls each port's switches appropriately to transfer the required charge between the link and the enabled ports.
A separate set of working registers can be used in some embodiments to control converter operations. Any value requiring a ramped rate of change can apply the rate of change to the working registers.
The mode set for a port during a given power cycle can determine what factor will drive the port's power level. This can be, for example, power, current, conductance, or net power. In “net power” mode, the port's power level can be set by, e.g., the sum of other port's power settings. The mode of at least one port will most preferably be set to net power in order to source or sink the power set by the other ports. If two ports are set as net power, the two ports will share the available power.
A main control state machine and its associated processes can control the transfer of power and charge between ports, as seen in FIG. 3C. The state machine can be controlled in turn by the contents of registers. The state machine transfers the amount of energy set by the interface from designated input ports to the link reactance, and then transfers the appropriate amount of energy from the link to designated output ports.
The Reset/Initialize state occurs upon a power reset, when converter firmware will perform self-tests to verify that the converter is functioning correctly and then prepare to start the converter. If no faults are found, the state machine proceeds to the Wait_Restart state.
The Wait_Restart state can be used to delay the start of the converter upon power up or the restart of the converter when certain faults occur. If a fault occurs, a bleed resistor is preferably engaged. Certain faults, once cleared, will preferably have a delay before restarting normal converter operation. The next state will be Startup.
When the Startup state begins, there is no energy in the link. This state will put enough energy into the link to resonate the link to the operational voltage levels, which are preferably greater than the highest voltage of any input line pair.
When starting from an AC port, the firmware will wait until a zero voltage crossing occurs on a line pair of the AC port. The firmware will then wait until the voltage increases to about 40 volts, then turn on the switches of the line pair for a short duration. This will put energy into the link and start the link resonating. The peak resonant voltage must be greater than the AC line pair for the next cycle. After the first energy transfer, more small energy transfers can be made to the link as the link voltage passes through the line pair voltage, increasing the link's resonant voltage until the link's peak voltage is equal to or greater than the first input line pair voltage. At this point, a normal power cycle is ready to start and the state will change to Power Cycle Start upon detection of a zero current crossing in the link.
In the Power Cycle Start state, the amount of charge and energy that will be transferred to or from the link and each port is determined at the start of a power cycle. This state begins on a link zero current crossing detection, so the link current will be zero at the start of the state. The link voltage will preferably be equal or greater than the highest input voltage.
The input and output line pairs that are not disabled is preferably sorted by their differential voltages from the highest voltage to the lowest voltage, where outputs are defined as having a negative voltage with respect to the start of the current power cycle. If the power factor of the AC port is not unity, one of the two line pairs of the AC port will switch between input and output for a portion of a 60 Hz waveform.
If a DC port's mode is set to have constant current or constant power, the constant current or power levels are converted to equivalent conductance values and used to adjust the relevant port's settings appropriately. If the port's mode is set to net power, the port will transfer the sum of all the energy of all other ports not in net power mode.
MPPT (Maximum Power Point Tracking) mode preferably constantly adjusts the charge put into the Link from a photovoltaic array to maximize transferred energy. There will typically be a maximum current draw after which voltage begins to decrease, where the particular maximal current depends on the photovoltaic array's output characteristics. This maximal current corresponds to maximum power, beyond which point energy transfer will decline. To determine this maximal point, energy transfer can be monitored while conductance is adjusted until a local maximum is found. There can be some variations in the amount of energy delivered, but this will tend to maximize energy transfer.
The charge Q to be transferred to the link can be found as, e.g., the product of conductance G, voltage V, and link power cycle period T (i.e. Q=G*V*T). The energy E to be transferred is then simply the product of the voltage times the charge (E=V*Q=G*V2*T).
Since other port operation modes prescribe the energy to be transferred to or from the link, at least one port is most preferably in “net power” mode. This assures that at least one port is most preferably thus dependent on the energy in the link, rather than prescribing the same, so that the amount of energy put into the link equals the amount of energy taken out of the link.
The amount of energy that is put into the link by other modes is summed together to determine the energy transfer to or from ports operating in net power mode. A small amount of energy can in some cases be subtracted from this sum if extra energy is to be added to the link this cycle. If multiple ports are operating in net power mode, the available energy is preferably split between the two ports according to, e.g., the Modbus registers. The amount of charge to be transferred is preferably determined by the relationship charge=energy/voltage.
For an AC port, the phase angle between the voltage and current on the AC port can be varied, based on e.g. power factor settings. An AC port can also source reactive current for AC port filter capacitors to prevent the filter capacitors from causing a phase shift.
Three-phase charge calculations for a three-phase AC port can, in some embodiments, proceed as follows. Zero crossing of the AC voltage waveform for a first phase is detected when the voltage changes from a negative to positive. This can be defined as zero degrees, and a phase angle timer is reset by this zero crossing. The phase angle timer is preferably scaled by the measured period of the AC voltage to derive the instantaneous phase angle between the voltage of this first phase and the zero crossing. The instantaneous phase angle can then be used to read the appropriate sinusoidal scalar from a sinusoidal table for the first phase. The instantaneous phase angle can then be adjusted appropriately to determine the sinusoidal scalars for the second and third phases.
The instantaneous phase angle of the first phase can be decremented by e.g. 90° to read a reactive sinusoidal scalar for the first phase, and then adjusted again to determine reactive sinusoidal scalars for the other two phases.
The required RMS line current of the port can then be determined, but can differ dependent on, e.g., whether the port is in net power mode is controlled by conductance. In conductance mode, RMS line current can be found by, e.g., multiplying the conductance for the AC port by its RMS voltage.
In net power mode, RMS line current can be found e.g. as follows. The energy transferred to the link by all ports not in net power mode is first summed to determine the net power energy available. The small amount of energy defined by the link energy management algorithm can be subtracted from the available energy if relevant. The net energy available is multiplied by the percentage of total power to be allocated to the present port, which is 100% if only one port is in net power mode: Power=Σ Energy*port %.
Line RMS current can then be found by dividing the energy for the AC port by the RMS voltage of the port, the link power cycle period, and square root of 3: line currentrms=Power/(timelink cycle*voltagerms*√3).
The instantaneous in-phase current can then be calculated, and will again differ based on the operational mode of the port. In a conductance mode, the three line-to-line instantaneous voltages can be multiplied by the port conductance to determine the instantaneous current of each phase.
In net power mode, the sinusoidal scalar for each phase can be multiplied by the RMS line current to determine the instantaneous current of each phase. Alternately, voltages from an analog/digital converter can be used to find the instantaneous currents directly: Instantaneous Current=energy*Va/d/(3*period*Vrms2). The charge can then be found as Q=energy*Va/d/(3*Vrms2).
RMS line reactive current can then be found e.g. from power factor as follows:

Power Factor=Power/(Power+reactive power)

reactive power=(Power/power factor)−Power

reactive powerline to line=Power/(3*power factor)−Power/3

rms reactive currentline=reactive powerline to line/rms voltageline to line.

Filter capacitive current can then be calculated from the filter capacitance values, line to line voltage, and frequency. Capacitive compensation current can then be added to the RMS line reactive current to determine the total RMS line reactive current. Total RMS reactive current can then be multiplied by the reactive sinusoidal scalar to derive the instantaneous reactive current for each phase.
The instantaneous current and the instantaneous current for each phase can then be added together and multiplied by the period of the link power cycle to determine the amount of charge to be transferred for each phase.
The energy to transfer to or from the link can be found by multiplying the charge value of each phase by the instantaneous voltage and summing the energy of the three phases together.
The phase with the largest charge will be dominant phase for this cycle, and the two line pairs for the AC port will be between the dominant phase and each of the other two phases. The amount of charge to be transferred for each line pair is preferably the amount of charge calculated for the non-dominant line of the pair. The next state will be the Charge Transfer state.
In the Charge Transfer state, a first line pair is selected and the respective switches turned on. Even though the switches are on, no conduction will occur until the voltage of the link drops below that of an input line pair, or rises above the voltage of an output line pair where appropriate. If one end of the link inductor reaches the voltage of one line of the line pair, that end of the link inductor is indirectly anchored to the respective line. The link inductor will subsequently not change in voltage until the respective switch is turned off.
The voltage of the line pair is then compared to the integrated link voltage. It is generally assumed that current will begin to flow through the switches once the integrated link voltage reaches the voltage of the line pair, minus a switch voltage drop. This switch voltage drop is assumed to be on the order of e.g. 8 V for a pair of switches.
The amount of charge flowing into or out of the link is monitored. The charge can be found as Q=ΣIΔt, or the sum of the current times the time interval.
The link current is typically approximately zero at the start of a power cycle. The link current increases through the end of the last input, then decreases until reaching zero at the beginning of the next power cycle. The link current can be found as I=Σ(VinstantaneousΔt/L), or the sum of the instantaneous voltage times the time interval divided by the inductance.
When the transferred charge is determined to have met the calculated amount for the given line pair, the state machine can progress to the next state. The next state can be Common Mode Management, or can be Idle. If the next state is Idle, all switches are turned off. In some sample embodiments, the state machine will only progress to the Common Mode Management state after the final output line pair.
The Common Mode Management state controls the common mode voltage of the link, as well as the energy left in the link following the prior state. To control the common mode voltage, one of the switches for the prior line pair is turned off, while the other switch is controlled by the Common Mode Management state. By having one switch on, the adjacent end of the link can be anchored at the respective line voltage. The voltage at the opposite end of the link can then increase until the current through the inductor drops to zero. The remaining switch can then be turned off. When a zero crossing is detected in the link current, the state machine will progress to the Idle state.
Two types of anchoring can be used in Common Mode Management. Direct anchoring occurs when one switch of a line pair is closed (turned on), which fixes the voltage of the nearest end of the link to the respective line voltage. While this switch is turned on, any change to the link's differential voltage will occur on the other end of the link, which will in turn change the link's common mode voltage.
Indirect anchoring occurs when both of a line pair's switches are turned on prior to a charge transfer. When the voltage of one end of the link is one switch-voltage-drop below the corresponding line voltage, the respective end of the link is anchored to that voltage. The voltage of the other end of the link will continue to change until the voltage across the link is equal to two switch-voltage-drops below the line pair voltage. At this point, charge transfer between the link and the line pair begins.
The Common Mode Management state also controls the energy left in the link after output charge transfer is completed, or after ramp-up. After the last output charge transfer, enough energy will most preferably remain in the link to have completed the last output charge transfer, and to cause the link voltages first to span, and then to decrease to just below, the voltages of the first input line pair. This can permit zero-voltage switching of the input switches. Zero-voltage switching, in turn, can reduce switching losses and switch overstressing. The voltages across the switches when conduction begins can preferably be e.g. 4 V, but is most preferably no more than 20 V. If insufficient energy remains in the link to permit zero-voltage switching, a small amount of power can be transferred from one or more ports in net power mode to the link during the subsequent power cycle.
FIG. 3D shows a sample embodiment in which the voltages of the last output span the voltages of the first input. It can be seen that the link-energy requirements have been met, though small amounts of energy can occasionally be needed to account for link losses.
FIG. 3E shows another sample embodiment in which the voltages of the last output are spanned by the voltages of the first input. Enough energy must be maintained in the link to resonate the link voltages to above the voltages of the first input. Additional energy can sometimes be needed to account for small link losses, but the link-energy requirements can be met fairly easily.
FIG. 3F shows a third sample embodiment, in which the voltages of the last output neither span nor are spanned by the voltages of the first input. Since the last output voltages do not span the first input voltages, the link voltage will need to be increased. Enough energy in the link needs to be maintained in the link to resonate the link voltages above the voltages of the first input pair before the link current crosses zero. This can in some sample embodiments require small amounts of additional energy to fulfill this requirement.
In each of the sample embodiments of FIGS. 3D-3F, the common mode voltage of the link will preferably be forced toward the common mode voltage of the first input. The switch of the last output furthest in voltage from the common mode voltage will preferably be turned off first. The link will thus first anchor to the end with a voltage closest to that desired while the other end changes. The other switch is preferably turned off either once the common mode voltage of the first input is turned off or else a zero-crossing is detected in the link current.
The Idle State most preferably ensures that all link switches remain for a period of time immediately after a switch is turned off. As switches do not turn off instantaneously, this can be used to minimize cross-conduction between lines, which can occur when one switch is turned on before another has time to completely turn off. In some sample embodiments in which the switches comprise e.g. IGBTs, the time between nominal and actual turn-off of the switches can be significant. After the requisite time has elapsed, the state machine can advance to the next state. If the prior state was the last line pair, the next state is preferably the Power Cycle Start state, and is otherwise preferably the Charge Transfer state.
In one sample embodiment, the bidirectional switches can comprise, e.g., two series IGBTs and two parallel diodes, as in FIG. 3G. In an embodiment like that of FIG. 3G, a bidirectional switch can have two control signals, each controlling one direction of current flow. Other bidirectional switches are also possible.
Switch control signals are most preferably monitored to prevent combinations of switches being turned which can lead to catastrophic failures of the converter. Only switches corresponding to a single line pair will preferably be enabled at a time. As relatively few possible switch combinations will prevent catastrophic failure, monitoring can look for the few permissible combinations to allow instead of looking for the many combinations to forbid.
Switch control signals can preferably also be monitored to avoid turning new switches on too quickly after another switch has been turned off. The switches take a finite time to turn off, and turning on another switch too quickly can cause damaging cross-conduction.
Voltage across each switch is also preferably monitored before it is turned on to avoid damaging overvoltage.
Zero-crossings in the link current are preferably detected e.g. using a toroid installed on a link cable. Instead of directly measuring link current, it can be calculated by integrating the voltage across the link and scaling the result. This calculated current can preferably be reset every time a zero-crossing is detected, to prevent long-term accumulation of error. Zero-crossings, when detected, can also be used to set the link polarity flag, as the voltage across the link reverses when the direction of current flow changes.
In some sample embodiments, power converter voltages can be measured with high-speed serial analog-to-digital (A/D) converters. In one sample embodiment, these converters can have e.g. a 3 MSPS (mega-samples per second) conversion rate. In one sample embodiment, the converters can take e.g. 14 clocks to start a conversion and clock in the serial data, leading to e.g. a data latency of 0.3 μs. One sample embodiment can use e.g. 22 such A/D converters.
Islanding occurs when a converter continues to output power when the AC power grid goes down. This can be extremely dangerous, especially for line crews attempting to fix the AC power grid. Islanding conditions are most preferably detected and used to trigger a shutdown of the converter's AC output.
Preferably ground fault detection is used on the DC inputs. When DC contactors are closed, the voltage drop between the common connection of a port's connectors and the DC port's ground connection will preferably be measured. If this voltage is over a certain limit, either too much ground current is present or else the port's ground fuse is blown. Both of these situations will generate a fault.
A fault will preferably be generated if toroids on input cables detect surges.
Each DC port will preferably have a pair of contactors connecting positive and negative power sources to an input ground connection. Configuration information is preferably read from the registers and used to open or close the contactors as needed. Before contactors are closed, DC filter capacitors are preferably pre-charged to the voltage on the line side of the contactors in order to prevent high-current surges across the contacts of the contactors.
An LCD or other type of screen is preferably provided as an interface to a power converter.
The temperature of a heat sink is preferably monitored and used to direct fans. Tachometers on the fans can preferably be monitored, and the information used to shut down fan control lines if a fan fails. As these temperature sensors can occasionally give incorrect information, in some sample embodiments e.g. two preceding readings can be compared against the current temperature reading, and e.g. the median value can be chosen as the current valid temperature.
In some sample embodiments, a processor can be used to control a power converter. This can be e.g. a NIOS processor which is instantiated in the field-programmable gate array.
In some sample embodiments, an interface to e.g. a 1 GB flash RAM can be used. In one sample embodiment, a flash RAM can have e.g. a 16-bit-wide bus and e.g. a 25-bit address bus. In some sample embodiments, an active serial memory interface can permit reading from, writing to, or erasing data from a serial configuration flash memory.
In some sample embodiments, a field-programmable gate array can be connected to e.g. a 1 MB serial nvSRAM with real time clock.
In some sample embodiments, dual row headers on a pc board can be used e.g. for testing and debugging purposes.
In some sample embodiments, LEDs or other indicators can be present on a control board. These indicators can be used e.g. for diagnostic purposes.
To minimize risks of condensation or other types of moisture damaging electronics, a power converter can preferably be kept in a sealed compartment. Some air flow is often necessary, however, due to e.g. temperature changes over time. Any air flowing into or out of the converter most preferably passes through one or more dehumidifiers. If left alone, the dehumidifiers eventually saturate and become useless or worse. Instead, heating elements can preferably be included with dehumidifiers to drive out accumulated moisture. When air flows into the otherwise-sealed compartment, dehumidifiers can remove moisture. When air flows out of the compartment, the heating elements can activate, so that ejected moisture is carried away with the outflowing air instead of continuing into the converter.
FIGS. 3DD and 3EE show two sample embodiments of bi-directional multi-port power conversion systems. In this sample embodiment, first input port 102 can include a power generator 202 connected to wind turbines 204, second input port 104 can include DC port for energy storage, and output port 108 can include an AC power grid.
According to one sample embodiment, generator 202 connected to wind turbines 204 can produce asynchronous AC, this asynchronous AC from generator 202 can be transformed to synchronous AC by power conversion module 106, and subsequently stored in second input port 104.
The present application discloses new approaches to power conversion. In particular, the inventor has developed various systems, devices and methods for achieving power conversion by using multiple soft-switched power modules connected in parallel.
Some exemplary parameters will be given to illustrate the relations between these and other parameters. However it will be understood by a person of ordinary skill in the art that these values are merely illustrative, and will be modified by scaling of further device generations, and will be further modified to adapt to different materials or architectures if used.
These techniques are applicable to a vast number of applications, including but not limited to all DC-DC, DC-AC, and AC-AC power conversions.
The present application discloses power converters which are generally of the Buck-Boost family, but which use capacitance, either parasitic alone or with added discrete device(s), in parallel with the Buck-Boost inductor to achieve low turn-off switching stresses (i.e. “soft switching”) on the semiconductor switches, allowing relatively slow and inexpensive switches to be used.
The innovative converter circuits, in various elements are constructed of input and output filter capacitances, and multiple power modules, ones of said power modules comprising: semiconductor switches, an inductor, and advantageously a capacitor in parallel with the inductor. Individual power modules can be independently controlled by a control means, which, controlling the input switches of an individual power module, first connects the inductor, initially at zero current, to the input voltage, which can be DC or the highest line-to-line voltage AC pair in a three phase input, except at startup, in which case a near zero-voltage line pair is used. The control then turns off those switches when the current reaches a point determined by the control to result in the desired rate of power transfer. The current then circulates between the inductor and capacitor, which results in a relatively low rate of voltage increase, such that the switches are substantially off before the voltage across them has risen significantly, resulting in low turn-off losses.
With DC or single phase AC input, no further current is drawn from the input. With 3 phase AC input, the control will again connect the inductor to the input lines, but this time to the line-to-line pair which has a lower voltage then the first pair. Turn on is accomplished as the relevant switches transition from reverse to forward bias. After drawing the appropriate amount of charge (which can be zero if the control determines that no current is to be drawn from the pair, as for example when the pair is at zero volts and input unity power factor is desired), the relevant switches are again turned off Under most conditions, the voltage on the inductor will then reverse (with relatively low rates of voltage change due to the parallel capacitance).
With 3 phase AC output, the control will turn on switches to allow current to flow from the inductor to the lowest voltage pair of lines which need current after the relevant switches become forward biased. The control turns off the switches after the appropriate amount of charge has been transferred. The inductor voltage then ramps up to the highest output line-to-line pair for 3 phase AC, or to the output voltage for single phase AC or DC.
Again, switches are turned on to transfer energy (charge) to the output, transitioning from reverse to forward bias as the voltage ramps up. If the output voltage is larger than the highest input voltage, the current is allowed to drop to zero, which turns off the switch with a low rate of current reduction, which allows for the use of relatively slow reverse recovery characteristics.
If the output voltage is less than the highest input voltage, the switches are turned off before current stops, so that the inductor voltage ramps up to the input voltage, such that zero-voltage turn on is maintained. Alternatively, the switches can be turned off before this point, so as to limit the amount of current into the output. In this case, the excess energy due to current in the inductor is directed back into the input by turning on switches to direct current flow from the inductor into either the highest voltage pair in three phase, or the single phase AC or DC input.
In a three phase AC converter, the relative charge per cycle allocated to each input and output line pair can be controlled to match the relative current levels on each line (phase). After the above scenario, when zero current is reached the inductor is reconnected to the input, but with a polarity reversed from the first connection, using switches that are complementary to the switches used in the first half of the cycle. This connection can occur immediately after zero current (or shortly after zero current if the input voltage is less than the output voltage, to allow the capacitor voltage time to ramp back down), giving full utilization of the power transfer capability of the inductor.
The full filter between the converter and an attached voltage source (utility) or sink (motor, another utility, or load) includes the line capacitance (line-to-line or line-to-neutral, as in Y or Delta), and a series line inductance (or line reactor as it's generally called). When driving a motor, the line reactance is the inductance of the motor. The full filter is a power filter, AND it does important conditioning for the converter.
Preferred sample converter embodiments benefit from having very low impedance voltage sources and sinks at the inputs and outputs. The link inductor current must be able to be very rapidly switched between the link capacitor and the I/O capacitors. Line reactance can prevent that from occurring, and can destroy the switches. The physical construction of the converter will preferably be carefully done to minimize all such inductance which can impair link reactance switching.
For proper operation, line capacitance will preferably be chosen so that the change in voltage on the line capacitance while charging or discharging the link inductance (e.g., ripple voltage) is only be a small fraction of the initial voltage, e.g., less than 10%. Line capacitance will preferably also be chosen so that the ripple current is within the capacitor's current rating.
Another important consideration is the resonant frequency formed by the L-C of the line reactance and the line capacitance (the I/O power filter). This frequency must be lower than the link power cycle frequency in order to not have that filter resonant with the voltage ripple on the line capacitance.
The capacitance needs to be large enough to reasonably stabilize the I/O voltage to allow the link inductor charge/discharge to occur properly, and the L-C resonant frequency needs to be less than twice the link voltage frequency, and generally at least 4 to 10 times lower.
It will also be noted that too much capacitance on line filter can lead to excess reactive power on the utility connection.
The present application teaches inter alia innovative converter circuits, which can be constructed of input and output filter capacitances, and multiple power modules, ones of said power modules comprising semiconductor switches, an inductor, and advantageously a capacitor in parallel with the inductor.
FIG. 6 shows a schematic of an example three phase converter 500 as shown in U.S. Pat. No. 7,599,196. FIG. 6 comprises an example embodiment of a power converter using a single power module functioning as described above. The innovative embodiments disclosed herein and further explained with respect to FIGS. 1-4 comprise multiple power modules, each of which function (and transfer energy) as described above and/or with respect to FIG. 6, connected in parallel between first and second (e.g., input and output) power portals.
As shown in FIG. 6, the converter 500 is connected to first and second power portals 522 and 523, each of which can source or sink power, and each with a port for each phase of the portal. It is the function of said converter 500 to transfer electric power between said portals while accommodating a wide range of voltage, current level, power factor, and frequency differences between the portals. Said first portal can be, for example, a 460 VAC three phase utility connection, while said second portal can be a three phase induction motor which is to be operated at variable frequency and voltage so as to achieve variable speed motor operation.
Referring to FIG. 6, the converter 500 comprises a first set of electronic switches S.sub.1u, S.sub.2u, S.sub.3u, S.sub.4u, S.sub.5u, and S.sub.6u that are connected between a first port 513 of a link inductor 520 and each phase, 524 through 529, of the input portal, and a second set of electronic switches S.sub.11, S.sub.21, S.sub.31, S.sub.41, S.sub.51, and S.sub.61 that are similarly connected between a second port 514 of link inductor 520 and each phase of the output portal. A link capacitor 521 is connected in parallel with the link inductor 520, forming the link reactance. Each of these switches is capable of conducting current and blocking current in both directions, and can be composed of, e.g., bi-directional IGBTs. Many other such bi-directional switch combinations are possible, such as e.g. anti-parallel reverse blocking IGBTs.
Generally, these switch combinations contain two independently controlled gates, with each gate controlling current flow in one direction. It is assumed herein, in describing the power modules comprising inventive embodiments, that two gate switches are used in each switch, and that the only gate enabled in a switch is the gate which controls current in the direction which is desired in the subsequent operation of the switch. Thus, when each switch mentioned herein is said to be enabled, said enabling occurs before conduction occurs, since that portion of the switch is reversed biased at the instant of being enabled, and does not conduct until it becomes forward biased as a result of the changing voltage on the parallel pair of inductor and capacitor. Switch embodiments which have only one gate, such as a one-way switch embedded within a full wage bridge rectifier, must be enabled only when the voltage across the switch is very small, which generally requires precise and accurate timing that can be difficult to achieve in practice.
Converter 500 also has input and output capacitor filters 530 and 531, respectively, which smooth the current pulses produced by switching current into and out of inductor 520. Optionally, a line reactor 532 can be added to the input to isolate the voltage ripple on input capacitor filter 531 from the utility and other equipment that can be attached to the utility lines. Similarly, another line reactor, not shown, can be used on the output if required by the application.
FIG. 1 shows an inventive embodiment with dual parallel power modules 100, which can be used to convert energy from input portal 102 passing through power modules 104 to output portal 106, while adjusting a wide range of voltages, current levels, power factors, and frequencies between portals. Each power module 104 can be constructed of 12 bi-directional switches 118 and a parallel inductor/capacitor (e.g., link 108). Bi-directional switches 118 represent sets of switches, different individual ones of said bi-directional switches 118 being connected to different ones of the phases of a corresponding power portal 102 or 106. Each bidirectional switch is capable of conducting and blocking current in two directions, and can be composed of bi-directional internal gate bipolar transistors (IGBTs) or other bidirectional switches. Most combinations of bidirectional switches contain two independently controlled gates, with each gate controlling current flow in one direction.
Each power module 104 includes link 108, which further includes link inductor 110 and link capacitor 112 connected in parallel, forming a resonant circuit that allows for soft switching and flexibility of adjusting link voltage to meet individual needs of input portal 102 and output portal 106.
Furthermore, input filter capacitors 114 can be placed between input phases, and output filter capacitors 114 can be placed between output phases, to closely approximate input portal 102 and output portal 106 and to attenuate current pulses produced by the bidirectional switches and link inductors 110. An input line reactor 116 can be needed in some applications to isolate the voltage ripple on the input filter capacitors 114. A line reactor 116 can also be used on the output side (not shown in FIG. 1) if required by the application.
Dual power modules 104 can be used whenever converter drive requirements exceed what is obtainable from a single power module 104 and/or when redundancy is desired for reliability reasons and/or to reduce I/O filter size, in addition to reducing costs and losses, and/or to increase available bandwidth. In addition, power modules 104 can be operated in a manner similar to multi-phase DC power supplies, such that link 108 reactance frequencies can be identical and current pulses drawn and supplied to input and output filter capacitors 114 from each module can be uniformly spaced in time. This operation mode can provide a more uniform current draw and supply, which can greatly reduce the per unit filtering requirement for a power converter.
FIG. 2 illustrates dual link current waveform 200 from dual parallel power modules 100. First link current 204 is from one power module 104, while second link current 206 is from the second power module 104.
First link current 204 and second link current 206 can operate 90 degrees out-of-phase 202, relative to the link inductors/capacitors 110, 112. 90 degrees out-of-phase 202 operation can produce a similar root mean square (RMS) current in the I/O filter capacitors 114 as single power module operation, while doubling the ripple frequency measured at the I/O filter capacitors.
Dual parallel power modules 100 can thus allow the same I/O filter capacitors 114 to be used for twice the total power, reducing the per unit I/O filter capacitance by a factor of about 2. In addition, since the ripple voltage is reduced by a factor of about 2, and the ripple frequency is doubled, the input line reactance requirement can be reduced by a factor of about 4. This can allow the total line reactor 116 mass to drop by a factor of about 2, reducing per unit line reactance requirement by a factor of about 4.
FIG. 4A shows a sawtooth waveform 300 from output filter capacitor 114 according to one sample embodiment of the present inventions. First link current 204 from one power module 104 shows one power module operating alone, leaving spaces between each first link current 204. Average current 302 flowing through output filter capacitor 114 can be 1⁄4 of first link current 204 pick (1⁄4 of current maximum). Note that link 108 reactances are only connected to each portal for half of each cycle, and that for each link 108 reactance, the period of connection to the first power portal 102 is mutually exclusive with the period of connection to the second power portal 106.
FIG. 4B shows an example of a sawtooth waveform 300 from output filter capacitor 114 with dual parallel power modules 100, showing that a second link current 206 from a second power module 104 can fill in the spaces between each first link current 204 sawtooth to produce a continuous current flow.
Current flowing out of filter capacitors 114 is generally equal to current flowing into filter capacitors 114. This can be accomplished by bi-directional switches 118 alternately connecting link 108 reactances to the first and second power portals 102, 106. Each power portal 102, 106 is substantially always connected to a different one of the power modules 104.
At the end of each half cycle, the connections flip, so that the two power modules 104 are separately connected to a different one of the power portals 102, 106 during each half cycle. Thus, the power modules 104 are substantially always, substantially simultaneously, connected to a different one of the power portals 102, 106 (and generally not to the same power portal 102 or 106), while also separately maintaining normal power module 104 operation as described hereinabove.
FIG. 5 shows an example of an output sawtooth waveform 300 from dual parallel power module 100. As shown, the average current 302 produced by a single power module can be doubled by adding a second power module, because the second link current 206 can fills in the spaces between each first link current 204 sawtooth. Average current 302 from dual parallel power module 100 can be above average current 302 from a single power module 104 (shown in FIG. 4A); thus, twice the total power can be achieved over the same amount of I/O filter capacitor 114.
In some embodiments, more than two power modules 104 can be used for additional options in multi-way conversion. In such cases, few modifications can be required for total power, current and voltage management, for topologies including more than two power modules 104. Higher power and better performance can be achieved through the use of multiple power modules 104 and one I/O filter capacitor 114.
According to some but not necessarily all embodiments, there is provided: A multiple power module soft-switched converter, comprising: first and second voltage source power portals, each with two or more ports by which electrical power is input or output to said portals; multiple power modules connected at least partially in parallel between said input portal and said output portal, ones of said power modules comprising: an inductive energy-transfer reactance, comprising paralleled capacitive and inductive components connected across a pair of busses, which operates in a full cycle mode with two power transfers for each cycle; and first and second full-bridge switch arrays, both connected across said pair of busses, each connected to a respective one of said power portals, and each comprising at least one bi-directional conducting and blocking switching device, which can block voltage and conduct current in both directions, for each said port of each said respective power portal; and a controller which is connected to each of said power modules respectively to operate said respective bridge switch arrays, with mutually exclusive conduction intervals between respective first and second ones of said bridge switch arrays, to drive current from both of said respective busses through said respective reactance in both directions during each cycle, and to commonly control all of said bridge switch arrays to minimize harmonics in the current drawn from and delivered to said input and output portals.
According to some but not necessarily all embodiments, there is provided: A multiple power module soft-switched converter, comprising: an input polyphase connection and an output polyphase connection; multiple capacitative components, ones of said capacitative components connected among the phases of and configured to filter said input polyphase connection, and ones of said capacitative components connected among the phases of and configured to filter said output polyphase connection; and a plurality N of power modules connected at least partially in parallel between said input polyphase connection and said output polyphase connection, ones of said power modules comprising: an energy-transfer reactance, comprising at least one inductor which has at least two terminals; an input switch array comprising multiple input switch pairs, each input switch pair connected to one phase of said input polyphase connection, and each input switch pair configured to selectably source or sink current to each of said terminals of said inductor at various different times; an output switch array connected to extract energy from said reactance when said input switch array is not active; wherein each of said input switch arrays performs at least two drive operations, in the same direction but from different ones of said phases, during a single half-cycle of said reactance, and wherein said power modules are configured to operate at inductor phase angles separated by 180/N degrees, whereby the capacitance of said capacitative components can be reduced, without increasing voltage ripple, compared to when N equals 1.
According to some but not necessarily all embodiments, there is provided: A dual power module soft-switched converter, comprising: first and second voltage source power portals, each with two or more ports by which electrical power is input or output to said portals; two power modules connected in parallel between said input portal and said output portal, ones of said power modules comprising: an inductive energy-transfer reactance, comprising paralleled capacitive and inductive components connected across a pair of busses, which operates in a full cycle mode with two power transfers for each cycle; and first and second full-bridge switch arrays, both connected across said pair of busses, each connected to a respective one of said power portals, and each comprising at least one bi-directional conducting and blocking switching device, which can block voltage and conduct current in both directions, for each said port of each said respective power portal; and a controller which is connected to respective ones of said power modules to operate said bridge switch arrays, with mutually exclusive conduction intervals between respective first and second ones of said bridge switch arrays, to drive current from both of said respective busses through said respective reactance in both directions during each cycle, and with substantially overlapping conduction intervals between each of said first bridge switch arrays and each non-respective one of said second bridge switch arrays, to drive current substantially continuously to output.
According to some but not necessarily all embodiments, there is provided: A method of operating a multiple power module soft-switched converter, comprising the actions of: a) operating a plurality N of first bridge switch arrays, ones of said first bridge switch arrays comprising bi-directional conducting and blocking switching devices, to operatively connect at least one terminal of a corresponding one of a plurality N of energy-transfer reactances to a power input, with polarity which reverses at different times, said energy-transfer reactances comprising paralleled capacitive and inductive components, which operate in a full cycle mode to output two power transfers for each cycle from each of said first bridge switch arrays; and b) operating a plurality N of second bridge switch arrays, ones of said second bridge switch arrays comprising bi-directional conducting and blocking switching devices, to operatively connect at least one terminal of a corresponding one of said energy-transfer reactances to a power output, with polarity which reverses at different times, said energy-transfer reactances operating in a full cycle mode to output two power transfers for each cycle from each of said second bridge switch arrays; wherein each of said reactances corresponds to a different pair of said first and second bridge switch arrays; wherein said pairs are connected at least partially in parallel with each other; and wherein said actions (a) and (b) are not performed simultaneously by ones of said pairs, and are performed to minimize harmonics in the current drawn and delivered by said power transfers.
According to some but not necessarily all embodiments, there is provided: A method of operating a multiple power module soft-switched converter, comprising: a) operating a plurality N of first bridge switch arrays, ones of said first bridge switch arrays comprising bi-directional conducting and blocking switching devices, to operatively connect at least one terminal of a corresponding one of a plurality N of energy-transfer reactances to a power input, with polarity which reverses at different times, said energy-transfer reactances comprising paralleled capacitive and inductive components, which operate in a full cycle mode to output two power transfers for each cycle from each of said first bridge switch arrays; b) operating a plurality N of second bridge switch arrays, ones of said second bridge switch arrays comprising bi-directional conducting and blocking switching devices, to operatively connect at least one terminal of a corresponding one of said energy-transfer reactances to a power output, with polarity which reverses at different times, said energy-transfer reactances operating in a full cycle mode to output two power transfers for each cycle from each of said second bridge switch arrays; and c) filtering said power input and said power output using capacitative components; wherein each of said reactances corresponds to a different pair of said first and second bridge switch arrays; wherein said pairs are connected at least partially in parallel with each other; wherein said actions (a) and (b) are not performed simultaneously by ones of said pairs, and are performed to operate said reactances at inductor phase angles separated by 180/N degrees, whereby the capacitance of said capacitative components can be reduced, without increasing voltage ripple, compared to when N equals 1.
According to some but not necessarily all embodiments, there is provided: A method of operating a dual power module soft-switched converter, comprising the actions of: a) operating two first bridge switch arrays, ones of said first bridge switch arrays comprising bi-directional conducting and blocking switching devices, to operatively connect at least one terminal of a corresponding one of two energy-transfer reactances to a power input, with polarity which reverses at different times, said energy-transfer reactances comprising paralleled capacitive and inductive components, which operate in a full cycle mode to output two power transfers for each cycle from each of said first bridge switch arrays; and b) operating two second bridge switch arrays, ones of said second bridge switch arrays comprising bi-directional conducting and blocking switching devices, to operatively connect at least one terminal of a corresponding one of said energy-transfer reactances to a power output, with polarity which reverses at different times, said energy-transfer reactances operating in a full cycle mode to output two power transfers for each cycle from each of said second bridge switch arrays; wherein each of said reactances corresponds to a different pair of said first and second bridge switch arrays; wherein different ones of said pairs are connected at least partially in parallel with each other; and wherein said actions (a) and (b) are not performed simultaneously by ones of said pairs, and are performed substantially simultaneously for said first and second bridge switch arrays from different ones of said pairs.
According to some but not necessarily all embodiments, there is provided: Methods and systems for transforming electric power between two or more portals using multiple power modules. Any or all portals can be DC, single phase AC, or multi-phase AC. Individual power modules comprise a plurality of bi-directional conducting and blocking semiconductor switches, and an inductor and parallel capacitor (reactance). The switches alternately connect the reactance between said portals, such that energy is transferred into the inductor from one or more input portals and/or phases, then transferred out of the inductor to one or more output portals and/or phases, with said parallel capacitor facilitating “soft” turn-off, and with any excess inductor energy being returned to the input. Dual power modules can operate 90 degrees out of phase. This configuration allows use of the same I/O filter capacitors as with a single power module, while achieving twice the total power produced by the power converter, reducing ripple voltage and doubling ripple frequency.
MODIFICATIONS AND VARIATIONS
As will be recognized by those skilled in the art, the innovative concepts described in the present application can be modified and varied over a tremendous range of applications, and accordingly the scope of patented subject matter is not limited by any of the specific exemplary teachings given. It is intended to embrace all such alternatives, modifications and variations that fall within the spirit and broad scope of the appended claims.
In some embodiments, operation without link capacitance is possible, at the expense of higher forward turn-off switching losses.
In some embodiments, additional portals are accommodated by the same power converter, as can be desired to accommodate power transfer to and from other power sources and/or sinks.
Embodiments have been described above using 12 bi-directional switches; in some embodiments, a different number and configuration of switches and configuration of switches can be used per power module, as shown, e.g., in U.S. Pat. No. 7,599,196, in embodiments described therein as inventive.
In some embodiments, multiple power modules are configured partially, rather than fully, in parallel.
Embodiments described above show particular configurations of power modules, input/output portals, line capacitance, etc.; in some embodiments, these configurations can be varied as shown, e.g., in U.S. Pat. No. 7,599,196, in embodiments described therein as inventive.
Additional general background, which helps to show variations and implementations, may be found in the following publications, all of which are hereby incorporated by reference: U.S. Pat. No. 5,977,569; U.S. Pat. No. 6,400,127; and U.S. Pat. No. 6,993,107.
Additional general background, which helps to show variations and implementations, as well as some features which can be implemented synergistically with the inventions claimed below, may be found in the following US patent applications. All of these applications have at least some common ownership, copendency, and inventorship with the present application, and all of them, as well as any material directly or indirectly incorporated within them, are hereby incorporated by reference: U.S. Pat. No. 8,406,265, U.S. Pat. No. 8,400,800, U.S. Pat. No. 8,395,910, U.S. Pat. No. 8,391,033, U.S. Pat. No. 8,345,452, U.S. Pat. No. 8,300,426, U.S. Pat. No. 8,295,069, U.S. Pat. No. 7,778,045, U.S. Pat. No. 7,599,196, US 2012-0279567 A1, US 2012-0268975 A1, US 2012-0274138 A1, US 2013-0038129 A1, US 2012-0051100 A1; US Provisionals 61/765,098, 61/765,099, 61/765,100, 61/765,102, 61/765,104, 61/765,107, 61/765,110, 61/765,112, 61/765,114, 61/765,116, 61/765,118, 61/765,119, 61/765,122, 61/765,123, 61/765,126, 61/765,129, 61/765,131, 61/765,132, 61/765,137, 61/765,139, 61/765,144, 61/765,146 all filed Feb. 15, 2013; 61/778,648, 61/778,661, 61/778,680, 61/784,001 all filed Mar. 13, 2013; 61/814,993 filed Apr. 23, 2013; 61/817,012, 61/817,019, 61/817,092 filed Apr. 29, 2013; 61/838,578 filed Jun. 24, 2013; 61/841,618, 61/841,621, 61/841,624 all filed Jul. 1, 2013; 61/914,491 and 61/914,538 filed Dec. 11, 2013; 61/924,884 filed Jan. 8, 2014; 61/925,311 filed Jan. 9, 2014; 61/928,133 filed Jan. 16, 2014; 61/928,644 filed Jan. 17, 2014; 61/929,731 and 61/929,874 filed Jan. 21, 2014; 61/931,785 filed Jan. 27, 2014; 61/932,422 filed Jan. 28, 2014; and 61/933,442 filed Jan. 30, 2014; and all priority applications of any of the above thereof, each and every one of which is hereby incorporated by reference.
None of the description in the present application should be read as implying that any particular element, step, or function is an essential element which must be included in the claim scope: THE SCOPE OF PATENTED SUBJECT MATTER IS DEFINED ONLY BY THE ALLOWED CLAIMS. Moreover, none of these claims are intended to invoke paragraph six of 35 USC section 112 unless the exact words “means for” are followed by a participle.
The claims as filed are intended to be as comprehensive as possible, and NO subject matter is intentionally relinquished, dedicated, or abandoned.

## Claims

1. A multiple power module soft-switched converter, comprising:
first and second voltage source power portals, each with two or more ports by which electrical power is input or output to said portals;
multiple power modules connected at least partially in parallel between said input portal and said output portal, ones of said power modules comprising:
an inductive energy-transfer reactance, comprising paralleled capacitive and inductive components connected across a pair of busses, which operates in a full cycle mode with two power transfers for each cycle; and
first and second switch arrays, both connected across said pair of busses, each connected to a respective one of said power portals, and each comprising at least one bi-directional conducting and blocking switching device, which can block voltage and conduct current in both directions, for each said port of each said respective power portal; and

a controller which is connected to each of said power modules respectively to operate said respective switch arrays, with mutually exclusive conduction intervals between respective first and second ones of said switch arrays, to drive current from both of said respective busses through said respective reactance in both directions during each cycle, and to commonly control all of said switch arrays to minimize harmonics in the current drawn from and delivered to said input and output portals.

2. The converter of claim 1, wherein said switch arrays include full bridge arrays.

3. The converter of claim 1, wherein, within ones of said power modules, said switch arrays are symmetrically connected to said energy-transfer reactance.

4. The converter of claim 1, wherein ones of said power modules further comprise a third switch array, which is connected to said reactance in parallel with said first and second switch arrays.

5. The converter of claim 1, wherein at least one of said power portals is shunted by a capacitor which provides a low-impedance voltage thereat.

6. The converter of claim 1, wherein at least one of said inductive components is implemented by a transformer.

7. A multiple power module soft-switched converter, comprising:
an input polyphase connection and an output polyphase connection;
multiple capacitative components, ones of said capacitative components connected among the phases of and configured to filter said input polyphase connection, and ones of said capacitative components connected among the phases of and configured to filter said output polyphase connection; and
a plurality N of power modules connected at least partially in parallel between said input polyphase connection and said output polyphase connection, ones of said power modules comprising:
an energy-transfer reactance, comprising at least one inductor which has at least two terminals;
an input switch array comprising multiple input switch pairs, each input switch pair connected to one phase of said input polyphase connection, and each input switch pair configured to selectably source or sink current to each of said terminals of said inductor at various different times;
an output switch array connected to extract energy from said reactance when said input switch array is not active;

wherein each of said input switch arrays performs at least two drive operations, in the same direction but from different ones of said phases, during a single half-cycle of said reactance, and
wherein said power modules are configured to operate at inductor phase angles separated by 180/N degrees,
whereby the capacitance of said capacitative components can be reduced, without increasing voltage ripple, compared to when N equals 1.

8. The converter of claim 7, wherein said switch arrays include full bridge arrays.

9. The converter of claim 7, wherein, within ones of said power modules, said switch arrays are symmetrically connected to said energy-transfer reactance.

10. The converter of claim 7, wherein ones of said power modules further comprise a third switch array, which is connected to said energy-transfer reactance in parallel with said first and second switch arrays.

11. The converter of claim 7, wherein at least one of said inductors is implemented by a transformer.

12. A dual power module soft-switched converter, comprising:
first and second voltage source power portals, each with two or more ports by which electrical power is input or output to said portals;
two power modules connected in parallel between said input portal and said output portal, ones of said power modules comprising:
an inductive energy-transfer reactance, comprising paralleled capacitive and inductive components connected across a pair of busses, which operates in a full cycle mode with two power transfers for each cycle; and
first and second switch arrays, both connected across said pair of busses, each connected to a respective one of said power portals, and each comprising at least one bi-directional conducting and blocking switching device, which can block voltage and conduct current in both directions, for each said port of each said respective power portal; and

a controller which is connected to respective ones of said power modules to operate said switch arrays,
with mutually exclusive conduction intervals between respective first and second ones of said switch arrays, to drive current from both of said respective busses through said respective reactance in both directions during each cycle, and
with substantially overlapping conduction intervals between each of said first switch arrays and each non-respective one of said second switch arrays, to drive current substantially continuously to output.

13. The converter of claim 12, wherein said switch arrays include full bridge arrays.

14. The converter of claim 12, wherein, within ones of said power modules, said switch arrays are symmetrically connected to said energy-transfer reactance.

15. The converter of claim 12, wherein ones of said power modules further comprise a third switch array, which is connected to said reactance in parallel with said first and second switch arrays.

16. The converter of claim 12, wherein at least one of said power portals is shunted by a capacitor which provides a low-impedance voltage thereat.

17. The converter of claim 12, wherein at least one of said inductive components is implemented by a transformer.