# Patent

## Title
Coherent optical receiver for pilot-assisted data transmission

## Abstract

A coherent optical receiver for a data-frame format in which a data frame has two or more pilot-symbol blocks, each having a cyclic prefix or suffix, and one or more payload-symbol blocks, each of which is concatenated with at least one adjacent block without a guard interval between them. The receiver uses optical signals corresponding to the pilot-symbol blocks to perform data-frame synchronization, frequency-offset correction, and channel-estimation procedures, which are robust even in the presence of certain transmission impairments. The receiver applies block-overlap processing with a sliding window to recover the payload data in a manner that substantially cancels the adverse effects of inter-block interference caused by the absence of guard intervals in the payload portion of the data frame. In one embodiment, the receiver performs channel-estimation and block-overlap processing for polarization-division-multiplexed signals in a manner that enables a combined, intertwined application of channel-response-compensation and polarization-demultiplexing procedures.

## Background
CROSS-REFERENCE TO RELATED APPLICATIONS
The subject matter of this application is related to that of U.S. Patent Application Publication No. 2012/148255, which is incorporated herein by reference in its entirety.
BACKGROUND
1. Field of the Invention
The present invention relates to optical communication equipment and, more specifically but not exclusively, to signal processing in coherent optical transmission systems.
2. Description of the Related Art
This section introduces aspects that may help facilitate a better understanding of the invention(s). Accordingly, the statements of this section are to be read in this light and are not to be understood as admissions about what is in the prior art or what is not in the prior art.
The next-generation of optical communication systems is being designed for relatively high data-transmission rates, e.g., higher than about 100 Gbit/s per channel. At these rates, the effects of chromatic dispersion (CD) and polarization-mode dispersion (PMD) significantly degrade the transmission performance of optical fiber links. Since practical implementation of dispersion compensation in the optical domain is relatively expensive, various digital-signal-processing (DSP) techniques, such as orthogonal frequency division multiplexing (OFDM), are used to mitigate the adverse effects of CD and PMD on signal transmission. However, one problem with optical OFDM is that it requires relatively sophisticated digital signal processing not only at the receiver, but also at the transmitter. In addition, optical OFDM has a relatively high peak-to-average power ratio (PAPR), which adversely affects the hardware cost by imposing rather stringent constraints on modulation and power-amplifier nonlinearities.
SUMMARY
Disclosed herein are various embodiments of a coherent optical receiver for a data-frame format in which a data frame has two or more pilot-symbol blocks, each having a cyclic prefix or suffix, and one or more payload-symbol blocks, each of which is concatenated with at least one adjacent block without a guard interval between them. The receiver uses optical signals corresponding to the pilot-symbol blocks to perform data-frame synchronization, frequency-offset correction, and channel-estimation procedures, which are robust even in the presence of transmission impairments, such as chromatic dispersion (CD) and polarization-mode dispersion (PMD). The receiver applies block-overlap processing with a sliding window to recover the payload data in a manner that substantially cancels the adverse effects of inter-block interference caused by the absence of guard intervals in the payload portion of the data frame. In one embodiment, the receiver performs channel-estimation and block-overlap processing for polarization-division-multiplexed signals in a manner that enables a combined, intertwined application of channel-response-compensation and polarization-demultiplexing procedures (e.g., as shown in FIG. 3C) that advantageously saves processing time and/or power.
According to one embodiment, provided is an optical receiver having an optical-to-electrical converter that mixes a modulated optical signal with an optical reference signal to generate a plurality of mixed optical signals, and converts the plurality of mixed optical signals into a corresponding plurality of digital electrical signals. The plurality of digital electrical signals comprises a first in-phase digital signal and a first quadrature-phase digital signal that have a first set of signal samples corresponding to a data frame. The data frame comprises a first plurality of pilot-symbol blocks and a first set of one or more payload-symbol blocks. Each pilot-symbol block comprises a respective plurality of pilot symbols and a respective guard interval. Each payload-symbol block comprises a respective plurality of constellation symbols from a first sequence of constellation symbols. At least one payload-symbol block and another block are concatenated without a guard interval between them, wherein the other block is either a pilot-symbol block of the same data frame or of the next data frame or a payload-symbol block of the same data frame. The receiver further has a digital signal processor that processes the first set of signal samples to recover the data encoded in the first sequence of constellation symbols.
According to another embodiment, provided is an optical communication method having the steps of: mixing a modulated optical signal with an optical reference signal to generate a plurality of mixed optical signals; and converting the plurality of mixed optical signals into a corresponding plurality of digital electrical signals. The plurality of digital electrical signals comprises a first in-phase digital signal and a first quadrature-phase digital signal that have a first set of signal samples corresponding to a data frame. The data frame comprises a first plurality of pilot-symbol blocks and a first set of one or more payload-symbol blocks. Each pilot-symbol block comprises a respective plurality of pilot symbols and a respective guard interval. Each payload-symbol block comprises a respective plurality of constellation symbols from a first sequence of constellation symbols. At least one payload-symbol block and another block are concatenated without a guard interval between them, wherein the other block is either a pilot-symbol block or a payload-symbol block. The method further has the step of processing the first set of signal samples to recover the data encoded in the first sequence.



BRIEF DESCRIPTION OF THE DRAWINGS
Other aspects, features, and benefits of various embodiments of the invention will become more fully apparent, by way of example, from the following detailed description and the accompanying drawings, in which:
FIG. 1 shows a block diagram of an optical transmission system according to one embodiment of the invention;
FIGS. 2A-2C illustrate the operation of a digital signal processor that can be used in the transmitter of the optical transmission system shown in FIG. 1 according to one embodiment of the invention; and
FIGS. 3A-3C illustrate the operation of a digital signal processor that can be used in the receiver of the optical transmission system shown in FIG. 1 according to one embodiment of the invention.



DETAILED DESCRIPTION
OFDM transmission and single-carrier (SC) transmission with frequency-domain equalization (FDE) have similar digital-signal-processing (DSP) complexities. The main DSP difference between these two signal transmission techniques is that, in OFDM transmission, the inverse fast-Fourier-transform (IFFT) operation is performed at the transmitter side while, in SC-FDE transmission, it is performed at the receiver side. As a result, a transmitter in an SC-FDE transmission system may use a digital-signal processor of lesser processing power than that of a digital-signal processor used in an OFDM transmitter. In addition, an SC-FDE transmission system can advantageously be configured to have a smaller peak-to-average power ratio (PAPR) value than a comparably performing OFDM system, thereby relaxing the constraints on modulation and power-amplifier nonlinearities.
FIG. 1 shows a block diagram of a single-carrier optical transmission system 100 according to one embodiment of the invention. System 100 has an optical transmitter 110 and an optical receiver 190 connected via a fiber link 150. In one embodiment, fiber link 150 is an amplified fiber link having one or more optical amplifiers (not explicitly shown in FIG. 1).
Transmitter 110 receives two independent data streams 102 and 104 for transmission to receiver 190. A digital-signal processor 120 processes data streams 102 and 104 as further described below in reference to FIGS. 2A-2C to generate digital signals 1221-1224. Digital signals 1221-1224 undergo a digital-to-analog conversion in digital-to-analog converters (DACs) 1241-1244, respectively, to produce drive signals 1261-1264. Drive signals 1261 and 1262 are in-phase (I) and quadrature-phase (Q) drive signals, respectively, corresponding to data stream 102. Drive signals 1263 and 1264 are similar in-phase and quadrature-phase drive signals corresponding to data stream 104.
An optical IQ modulator 140X uses drive signals 1261 and 1262 to modulate an optical-carrier signal 132X generated by a laser source 130 and to produce a modulated signal 142X. An optical IQ modulator 140Y similarly uses drive signals 1263 and 1264 to modulate an optical-carrier signal 132Y generated by laser source 130 and to produce a modulated signal 142Y. A polarization beam combiner 146 combines modulated signals 142X and 142Y to produce an optical polarization-division-multiplexed (PDM) signal 148. Note that optical-carrier signals 132X and 132Y have the same carrier frequency. Each of drive signals 126 can be amplified by an RF amplifier (not explicitly shown) before being applied to drive the corresponding optical IQ modulator 140.
Fiber link 150 receives signal 148 from beam combiner 146 for transmission to receiver 190. While propagating through fiber link 150, signal 148 is subjected to various transmission impediments, such as chromatic dispersion (CD) and polarization mode dispersion (PMD), and emerges at the receiver end of the fiber link as an optical signal 152.
Receiver 190 has an optical-to-electrical (O/E) converter 160 having (i) two input ports labeled S and R and (ii) four output ports labeled 1 through 4. Input port S receives optical signal 152. Input port R receives an optical reference signal 158 generated by an optical local oscillator (OLO) 156. Reference signal 158 has substantially the same optical-carrier frequency (wavelength) as signal 152. Reference signal 158 can be generated, e.g., using a tunable laser controlled by a wavelength-control loop (not explicitly shown in FIG. 1) that forces an output wavelength of the tunable laser to substantially track the carrier wavelength of signal 152. In various embodiments, optical local oscillator 156 may comprise a combination of tunable and/or non-tunable lasers, optical frequency converters, optical modulators, and optical filters appropriately connected to one another to enable the generation of reference signal 158.
O/E converter 160 mixes input signal 152 and reference signal 158 to generate eight mixed optical signals (not explicitly shown in FIG. 1). O/E converter 160 then converts the eight mixed optical signals into four electrical signals 1621-1624 that are indicative of complex values corresponding to the two orthogonal-polarization components of signal 152. For example, electrical signals 1621 and 1622 may be an analog in-phase signal and an analog quadrature-phase signal, respectively, corresponding to an x-polarization component of signal 152. Electrical signals 1623 and 1624 may similarly be an analog in-phase signal and an analog quadrature-phase signal, respectively, corresponding to a y-polarization component of signal 152.
In one embodiment, O/E converter 160 is a polarization-diverse 90-degree optical hybrid (PDOH) with four balanced photo-detectors coupled to its eight output ports. Various suitable PDOHs are commercially available, e.g., from Optoplex Corporation of Fremont, Calif., and CeLight, Inc., of Silver Spring, Md. Additional information on various O/E converters that can be used to implement O/E converter 160 in various embodiments of system 100 are disclosed, e.g., in U.S. Patent Application Publication No. 2010/0158521, U.S. patent application Ser. No. 12/541,548 (filed on Aug. 14, 2009), and International Patent Application No. PCT/US09/37746 (filed on Mar. 20, 2009), all of which are incorporated herein by reference in their entirety.
Each of electrical signals 1621-1624 generated by O/E converter 160 are converted into digital form in a corresponding one of analog-to-digital converters (ADCs) 1661-1664. Optionally, each of electrical signals 1621-1624 may be amplified in a corresponding amplifier (not explicitly shown) prior to the resulting signal being converted into digital form. Digital signals 1681-1684 produced by ADCs 1661-1664 are processed by a digital signal processor 170, e.g., as further described below in reference to FIGS. 3A-3C, to recover the data applied by data streams 102 and 104 to transmitter 110. The recovered data are outputted from receiver 190 via output signals 192 and 194, respectively.
FIGS. 2A-2C illustrate the operation of a digital signal processor 200 that can be used to implement digital signal processor 120 (FIG. 1) according to one embodiment of the invention. More specifically, FIG. 2A shows a block diagram of processor 200. FIGS. 2B-2C show an exemplary frame structure used by digital signal processor 200.
Processor 200 processes an input data stream 202 to generate digital output signals 222I and 222Q. In a representative embodiment, processor 120 (FIG. 1) is implemented using two processors 200 configured to operate in parallel to one another. More specifically, for the first of these two parallel processors 200, input data stream 202 is data stream 102, and digital output signals 222I and 222Q are digital signals 1221 and 1222, respectively. For the second of these two parallel processors 200, input data stream 202 is data stream 104, and digital output signals 222I and 222Q are digital signals 1223 and 1224, respectively.
Input data stream 202 is applied to a coding module 204, where it is optionally interleaved and subjected to forward-error-correction (FEC) coding.
A coded bit stream 206 produced by coding module 204 is applied to a constellation-mapping module 208, where it is converted into a corresponding sequence 210 of constellation symbols. The constellation used by constellation-mapping module 208 can be, for example, a QAM (Quadrature Amplitude Modulation) constellation or a QPSK (Quadrature Phase Shift Keying) constellation.
Symbol sequence 210 is applied to a framing module 212, where it is converted into a corresponding sequence 214 of data frames. When processor 200 is used to implement DSP 120 (FIG. 1), sequence 214 consists of two parallel subsequences, one corresponding to the X polarization and the other corresponding to the Y polarization. Frame sequence 214 produced by framing module 212 is then applied to a pulse-shaping module 218, where it is converted into output signals 222I and 222Q.
FIG. 2B shows an exemplary structure of frame sequence 214 generated by framing module 212. A representative frame 230 of frame sequence 214 comprises a plurality of pilot-symbol blocks (PSs) and a plurality of payload-symbol blocks (DSs). For example, frame 230 has (i) pilot-symbol blocks PSs, PSc1, and PSc2 and (ii) payload-symbol blocks DS1, DS2, . . . , DSn. One important difference between a pilot-symbol block PS and a payload-symbol block DS is that the former has a cyclic prefix (CP, also sometimes referred to as a guard interval) while the latter does not. The use of cyclic prefixes in pilot-symbol blocks PS helps the synchronization and channel-estimation procedures performed at the receiver, e.g., receiver 190 (FIG. 1), to be robust in the presence of transmission impairments, such as CD and/or PMD. The nonuse of cyclic prefixes in payload-symbol blocks DS helps to minimize the transmission overhead and thus achieve relatively high payload-data throughput. Channel-compensation procedures applied to the CP-free payload-symbol blocks (blocks DS in FIG. 2B) rely on the channel information obtained from pilot-symbol blocks PSc1 and PSc2 and are described in more detail below in reference to FIGS. 3A-3C.
Pilot-symbol block PSs is designed to aid the receiver, e.g., receiver 190 (FIG. 1), in frequency estimation and frame synchronization. An exemplary pilot-symbol block PSs comprises (i) cyclic prefix CPs and (ii) symbol sequence E(n) having 2N symbols (i.e., n=1, 2, . . . 2N), in which the string that has the first N symbols is identical to the string that has the last N symbols. Cyclic prefix CPs has a copy of the last L symbols of symbol sequence E(n). In mathematical terms, pilot-symbol block PSs is expressed by Eq. (1):

PSS=[E(n=(2N−L+1):2N),E(n=1:2N)]  (1)

where the “:” symbol indicates a range for n starting from the value to the left of the symbol and ending with the value to the right of the symbol.

In one embodiment, symbol sequence E(n) is constructed as follows. First, a set of 2N orthogonal (in the OFDM sense) frequencies is selected and consecutively numbered starting from the lowest and ending with the highest frequency. Second, each of the odd-numbered frequencies is assigned the amplitude of zero, and each of the even-numbered frequencies is assigned a symbol that is randomly selected from a QPSK constellation. Recall that a QPSK constellation consists of four constellation points positioned, with uniform angular spacing, on a circle that is centered on the origin of a complex plane. The result of this assignment is a set having 2N frequency-domain symbols, half of which are zeros. Finally, an inverse fast-Fourier-transform (IFFT) operation is applied to this set of 2N frequency-domain symbols to arrive at symbol sequence E(n).
In another embodiment, symbol sequence E(n) is expressed by Eq. (2):

E(n)=exp(−jπ(n−1)2/N)  (2)

where n=1, 2, . . . , 2N.

By having two identical halves, symbol sequence E(n) can readily be used for autocorrelation to find the starting point of the frame, and to find the frequency offset between carrier signals 132 and reference signal 158. Suitable methods that can be used to perform autocorrelation-based synchronization are described, e.g., in an article by T. M. Schmidl and D. C. Cox entitled “Robust Frequency and Timing Synchronization for OFDM,” published in IEEE Transactions on Communications, Vol. 45, No. 12, December 1997, pp. 1613-1621, which article is incorporated herein by reference in its entirety.
In one embodiment, pilot-symbol block PSs corresponding to the X-polarization (e.g., represented by modulated signal 142X, FIG. 1) is the same as pilot-symbol block PSs corresponding to the Y-polarization (e.g., represented by modulated signal 142Y, FIG. 1). In an alternative embodiment, the X- and Y-polarizations may use different respective pilot-symbol blocks PSs.
FIG. 2C shows an exemplary structure of pilot-symbol blocks PSc1 and PSc2. Note that the pilot-symbol blocks PSc1 and PSc2 corresponding to the X-polarization may differ from the pilot-symbol blocks PSc1 and PSc2 corresponding to the Y-polarization. Pilot-symbol blocks PSc1 and PSc2 are designed to aid the receiver, e.g., receiver 190 (FIG. 1), in channel estimation and channel compensation.
Similar to pilot-symbol block PSs, each pilot-symbol block PSc has a cyclic prefix (CP) prepended to a respective (known, predetermined) body sequence of symbols (KS). Cyclic prefix CP is constructed by taking several symbols from the end of body sequence KS, which makes each pilot-data set PSc a partially cyclic sequence. The length (L) of cyclic prefix CP is selected to be longer than the expected duration of the impulse response of the channel, e.g., fiber link 150, FIG. 1. This property of pilot-symbol blocks in frame sequence 214 enables the intended receiver to process the pilot-symbol blocks in the received signal (e.g., signal 152, FIG. 1) in a manner that mitigates inter-block interference imposed by the fiber link due to the effects of CD and PMD.
Body sequences KS corresponding to pilot-symbol blocks PSc1 and PSc2 have the same length, which is larger than the length of a payload-symbol block DS in one embodiment (also see the description of FIG. 3C below). In a representative implementation, each body sequence KS is a special polyphase sequence of length 2N, wherein the amplitudes of all non-zero symbols in the time domain have different phases but the same amplitude, where N is a positive integer. One purpose of using this type of a sequence is to enable the receiver to accurately and uniformly probe the channel-response function, H, over the entire frequency range of interest. Channel-response function H is a frequency-dependent 2×2 matrix whose elements are complex functions of frequency that describe the combined signal-transfer characteristics of the front end of the transmitter (e.g., transmitter 110, FIG. 1), the fiber link (e.g., fiber link 150, FIG. 1), and the front end of the receiver (e.g., receiver 190, FIG. 1).
In one embodiment, each body sequence KS has 2N symbols, and the individual body sequences KS1-KS4 shown in FIG. 2C are expressed by Eqs. (3)-(5):

KS1(n)=exp(−jπ(n−1)2/N)  (3)

KS2(n)=exp(−jπn(n−1)/N)  (4)

KS3=KS2,KS4=KS1  (5)

where n=1, 2, . . . , 2N. The relationship between different body sequences KS expressed by Eq. (5) may be advantageous in that it enables the receiver to readily calculate, in the frequency domain, all four elements of the 2×2 channel-response matrix as a function of frequency. A suitable matrix-calculation method that can be used for this purpose is described, e.g., in an article by C. J. Youn, “An Efficient and Frequency-Offset-Tolerant Channel Estimation and Synchronization Method for PDM CO-OFDM Transmission,” published in the 2010 European Conference on Optical Communications (ECOC'10) as paper P4.06.

In alternative embodiments, additional suitable polyphase sequences can be constructed from the polyphase sequences defined by Eqs. (3)-(4), e.g., by one or more of the following: (i) cyclically shifting the entire sequence; (ii) phase shifting each symbol of the sequence by a constant phase; (iii) taking the m-th power of each symbol of the sequence, where m is an integer greater than one; and (iv) phase-conjugating the entire sequence. Additional suitable polyphase sequences (or codes) with uniform amplitudes in both the time and frequency domains can be constructed based on the general description provided in the article by David C. Chu, “Polyphase Codes with Good Periodic Correlation Properties,” published in IEEE Transactions on Information Theory, July 1972, pp. 531-532, which is incorporated herein by reference in its entirety.
Note that pilot-symbol blocks PSc1 and PSc2 corresponding to the X polarization are synchronous with pilot-symbol blocks PSc1 and PSc2, respectively, corresponding to the Y polarization. This characteristic of pilot-symbol blocks PSc1 and PSc2 is a manifestation of a more-general characteristic of data frame 230, according to which each symbol block corresponding to the X polarization is synchronous with a counterpart symbol block corresponding to the Y polarization. As such, pilot-symbol blocks PSs corresponding to the X polarization is synchronous with pilot-symbol block PSs corresponding to the Y polarization. Similarly, each payload-symbol block DSi corresponding to the X polarization is synchronous with pilot-symbol block DSi corresponding to the Y polarization (see FIG. 2B).
Referring back to FIG. 2A, frame sequence 214 produced by framing module 212 is applied to a pulse-shaping module 218, where it is converted into output signals 222I and 222Q. Pulse shaping implemented in pulse-shaping module 218 is a process of generating a digital waveform that, after being converted into a corresponding analog signal, can be applied to an optical modulator to modulate an optical-carrier signal, such as optical-carrier signal 132X or 132Y (FIG. 1), so that the resulting modulated optical signal is modulated with symbols corresponding to the various blocks of frame sequence 214. In one configuration, oversampling may be applied, e.g., by duplicating each signal sample one or more times for output signals 222I and 222Q.
FIGS. 3A-3C illustrate the operation of a digital signal processor 300 that can be used to implement digital signal processor 170 (FIG. 1) according to one embodiment of the invention. More specifically, FIGS. 3A-3B show block diagrams of processor 300. FIG. 3C shows the processing implemented in an FDCCE (frequency-domain-channel-compensation/equalization) sub-module 352 of processor 300. When processor 300 is used as processor 170, input signals 3021-3024 correspond to signals 1681-1684 (FIG. 1), respectively, and output signals 332x and 332y correspond to output signals 192 and 194 (FIG. 1), respectively.
EDC (electronic dispersion compensation) modules 310 perform digital signal processing that mitigates the detrimental effects of chromatic dispersion imposed on input signal 152 by fiber link 150. In particular, EDC module 310x processes input signals 3021-3022 corresponding to the first principal polarization axis (e.g., the x axis) of PDOH 160 (FIG. 1). Similarly, EDC module 310y processes input signals 3023-3024 corresponding to the second principal polarization axis (e.g., the y axis) of PDOH 160. Note that the X and Y polarizations used at the transmitter may or may not be aligned with the principal polarization axes (i.e., the x and y axes) of PDOH 160. Various EDC modules that can be used to implement EDC module 310 are disclosed, e.g., in U.S. Pat. Nos. 7,570,889, 7,532,820, and 7,382,984, all of which are incorporated herein by reference in their entirety.
Dispersion-compensated signals 3121-3124 produced by EDC modules 310x and 310y are applied to a pilot-assisted frequency-division-equalization and decoding (PA-FDED) module 320 that processes these signals, e.g., as further described below in reference to FIGS. 3B-3C, to recover the original data streams applied to the transmitter for transmission, e.g., data streams 102 and 104 of FIG. 1. More specifically, PA-FDED module 320 generates data stream 332x that carries the data corresponding to the first independently modulated component of signal 152 (e.g., component 142X, FIG. 1), and similarly generates data stream 332y that carries the data corresponding to the second independently modulated component of signal 152 (e.g., component 142Y, FIG. 1).
FIG. 3B shows a block diagram of PA-FDED module 320 according to one embodiment of the invention. PA-FDED module 320 has a synchronization sub-module 340 that receives, as its input, dispersion-compensated signals 3121-3124. Synchronization sub-module 340 relies on the above-described properties of pilot-symbol blocks PSs to determine the start of each frame 230 (also see FIG. 2B). In one configuration, synchronization sub-module 340 calculates an autocorrelation function defined by Eq. (6):




  
    
      
        
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where the “*” symbol denotes a complex conjugate; rx(n) is a complex value whose real part is a signal sample provided by signal 3121 and whose imaginary part is a corresponding signal sample provided by signal 3122; and ry(n) is a complex value whose real part is a signal sample provided by signal 3123 and whose imaginary part is a corresponding signal sample provided by signal 3124. Since symbol sequence E(n) of pilot-symbol block PSs has two identical portions of length N, the absolute value of function P(n) has a pronounced maximum that is temporally aligned with the first symbol of symbol sequence E(n) and, as such, can be used to determine the temporal position of the corresponding frame 230.

An FE (frequency-estimation/compensation) sub-module 344 that is located downstream from synchronization sub-module 340 performs electronic estimation and compensation of a mismatch between the carrier-frequency of input signal 152 and the frequency of reference signal 158 (see FIG. 1). In one embodiment, FE sub-module 344 determines the phase of P(nmax), where nmax is the time slot corresponding to the maximum of the absolute value of function P(n) (see Eq. (6)) determined by synchronization sub-module 340. FE sub-module 344 then uses the phase of P(nmax) to calculate the frequency offset between signals 152 and 158. If the frequency offset is zero, then P(nmax) is real and its phase is zero. If the frequency offset is not zero, then P(nmax) is complex and its phase is directly related to the frequency offset and the duration of pilot-symbol block PSs. Because the frequency offset may change over time, FE sub-module 344 performs the frequency-offset calculation for each received frame 230.
After the frequency offset is determined, FE sub-module 344 performs frequency-mismatch compensation by applying to each signal sample a phase shift equal to the frequency offset multiplied by 2π and the time elapsed between the start of the frame determined by synchronization sub-module 340 and the signal sample. Various FE modules that can be adapted to function as FE sub-module 344 are disclosed, e.g., in U.S. Pat. No. 7,747,177 and U.S. Patent Application Publication No. 2008/0152361, both of which are incorporated herein by reference in their entirety.
A CE (channel-estimation) sub-module 348 uses signal samples corresponding to pilot-symbol blocks PSc1 and PSc2 to determine the channel-response function, H, which can be expressed in the frequency domain as a 2×2 Jones matrix given by Eq. (7):




  
    
      
        
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More specifically, using the a priori knowledge of body sequences KS1, KS2, KS3, and KS4, CE sub-module 348 constructs channel-response function H so that the application of that channel-response function to these body sequences transforms them into the received signal samples corresponding to these known body sequences. Note that an individual channel-response function H can be calculated by CE sub-module 348 for each frame 230, which enables the receiver to appropriately track time-dependent channels.

FIG. 3C also shows exemplary processing implemented in FDCCE (frequency-domain channel-compensation/equalization) sub-module 352. More specifically, the shown processing handles signal samples corresponding to the payload of a single transmitted frame (e.g., payload-symbol blocks DS in frame 230 having data corresponding to both the X and Y polarizations, see FIG. 2B) and, as such, uses the channel-response function H determined by channel-estimation sub-module 348. For processing each new frame, FDCCE sub-module 352 uses a corresponding new channel-response function H determined by and received from CE sub-module 348.
As already indicated above, the symbols of payload-symbol blocks DS in frame 230 do not have cyclic prefixes. As a result, inter-block interference (IBI) occurs at the receiver. To mitigate the adverse effects of IBI, FDCCE sub-module 352 performs, as further described below and with respect to FIG. 3C, overlap FDCCE processing with a sliding window 368 that has 2N consecutive signal samples from a sequence 366 of frequency-corrected signal samples corresponding to the payload of a received frame. Sequence 366 consists of two parallel sub-sequences, i.e., a sequence 366x that corresponds to signals 3121-3122 and a sequence 366y that corresponds to signals 3123-3124 (also see FIGS. 3A-3B). Note that signal samples in sequence 366x have contributions both from the symbols carried by signal 142X and from the symbols carried by signal 142Y (FIG. 1). Signal samples in sequence 366Y similarly have contributions both from the symbols carried by signal 142X and from the symbols carried by signal 142Y (FIG. 1).
For an i-th position of sliding window 368, FDCCE sub-module 352 applies: (i) a fast Fourier transform (FFT) 372ix to a block 370ix of 2N signal samples located inside the window and belonging to sequence 366x and (ii) an FFT 372iy to a block 370iy of 2N signal samples located inside the window and belonging to sequence 366y. FFT operation 372ix converts block 370ix into a corresponding block 374ix of 2N frequency components. FFT operation 374 similarly converts block 370iy into a corresponding block 374iy of 2N frequency components.
Blocks 374ix and 374iy are subjected to a frequency-domain-equalization (FDE) procedure 376i, which converts these blocks into blocks 378ix and 378iy. Similar to blocks 374i, each of blocks 378ix and 378iy has 2N frequency components. FDE procedure 376i includes the application of inverse channel-response function H−1, which is derived by FDCCE sub-module 352 from channel-response function H (see Eq. (7)) determined by channel-estimation sub-module 348. Since channel-response function H generally has a non-diagonal form (i.e., b(f)≠0 and c(f)≠0), FDE procedure 376i mixes frequency components from blocks 374ix and 374iy to produce the corresponding frequency components for blocks 378ix and 378iy. Note that FFT operations 372i do not have this feature because (i) block 374ix is produced by operating solely on block 370ix and without using any signal samples from block 370iy and (ii) block 374iy is produced by operating solely on block 370iy and without using any signal samples from block 370ix.
Blocks 378ix and 378iy are subjected to inverse FFT (IFFT) operations 380ix and 380iy, respectively. IFFT operation 380ix converts block 378ix into 2N time-domain signal samples, which form block 382ix. IFFT operation 380iy similarly converts block 378iy into 2N time-domain signal samples, which form block 382iy. Similar to FFT operations 372i, IFFT operations 380i do not mix the x and y blocks.
Block 382ix is truncated to remove Ne signal samples from the beginning of the block and Ne signal samples from the end of the block, where Ne is a predetermined number chosen so that the signal samples affected by IBI are being truncated out. The remaining portion of block 382ix having 2N-2Ne signal samples is used to form sequence 384x of equalized signal samples that is output from FDCCE sub-module 352 to the downstream sub-modules of PA-FDED module 320. Block 382iy is similarly truncated to remove Ne signal samples from the beginning of the block and Ne signal samples from the end of the block. The remaining portion of block 382iy having 2N-2Ne signal samples is used to form sequence 384y of equalized signal samples that is also output from FDCCE sub-module 352 to the downstream sub-modules of PA-FDED module 320.
Sliding window 368 is shifted down sequences 366x and 366y by 2N-2Ne signal samples to the (i+1)-th position, and the processing that is described above for the i-th is repeated for the (i+1)-th position as indicated in FIG. 3C. The shifts of sliding window 368 by 2N-2Ne signal samples and the corresponding processing are repeated until all of the payload symbols of the corresponding frame 230 for both X- and Y-polarizations have been recovered.
In one configuration, Ne=L/2 and 2N-2Ne=NDS, where L is the length of cyclic prefix CP in pilot-symbol blocks PSc1 and PSc2 (FIG. 2C), and NDS is the length of a payload-symbol block (DS) in frame 230 (FIG. 2B). In this configuration, the processing performed at the i-th position of sliding window 368 recovers all symbols of payload-symbol block DSi in frame 230 (see FIG. 2B). To recover the symbols of the first payload-symbol block in the frame (DS1), sliding window 368 is positioned so that the first Ne signal samples inside the window belong to pilot-symbol block PSc2 of the same frame. To recover the symbols of the last payload-symbol block in the frame (DSn), sliding window 368 is positioned so that the last Ne signal samples inside the window belong to pilot-symbol block PSs of the next frame.
Note that, as illustrated above, CE sub-module 348 can perform channel-estimation and FDCCE sub-module 352 can perform block-overlap processing for polarization-division-multiplexed signals in a manner that enables FDCCE sub-module 352 to apply channel-response-compensation and polarization-demultiplexing procedures in a combined, intertwined, non-serial manner that advantageously saves processing time and/or power.
Referring back to FIG. 3B, equalized sequences 384x and 384y produced by FDCCE sub-module 352 are applied to a phase-estimation/phase-correction (PE/PC) sub-module 356. PE/PC sub-module 356 performs digital processing that corrects or compensates for slowly changing phase shifts between input signal 152 and reference signal 158 (FIG. 1), and then estimates the phase of the signal samples in equalized sequences 384x and 384y for constellation demapping and decoding. Various processing modules that can be used to implement PE/PC sub-module 356 are disclosed, e.g., in above-cited U.S. Patent Application Publication No. 2008/0152361 and also in U.S. Pat. No. 7,688,918 and U.S. Patent Application Publication No. 2008/0075472, both of which are incorporated herein by reference in their entirety.
A demapping sub-module 360 uses the phase estimates obtained by PE/PC sub-module 356 and the constellation map to convert equalized sequences 384x and 384y into the corresponding sequences of constellation symbols. Demapping sub-module 360 then decodes each constellation symbol to convert it into the corresponding set of bits, thereby generating bit streams 362x and 362y, corresponding to equalized sequences 384x and 384y, respectively. In the absence of errors, bit stream 362x is a copy of bit stream 206X, and bit stream 362y is a copy of bit stream 206Y (also see FIG. 2A). When decoding errors are present, a bit stream 362 may differ somewhat from the corresponding bit stream 206.
An FEC (forward error correction) sub-module 364 performs error correction in bit streams 362x and 362y using data redundancies that were introduced into the corresponding bit streams 206 by coding modules 204 (FIG. 2A). The resulting error-corrected bit streams are output via signals 332x and 332y. Many FEC methods suitable for use in FEC sub-module 364 are known in the art. Both hard-decision and soft-decision decoding may be implemented in various embodiments of FEC sub-module 364. Several representative examples of such methods are disclosed, e.g., in U.S. Pat. Nos. 7,734,191, 7,574,146, 7,424,651, 7,212,741, and 6,683,855, all of which are incorporated herein by reference in their entirety.
While this invention has been described with reference to illustrative embodiments, this description is not intended to be construed in a limiting sense. Although various embodiments of the invention have been described in reference to polarization-division-multiplexed (PDM) signals, the invention is not so limited and may be similarly applied to processing non-PDM signals. Various modifications of the described embodiments, as well as other embodiments of the invention, which are apparent to persons skilled in the art to which the invention pertains are deemed to lie within the principle and scope of the invention as expressed in the following claims.
The term “single-carrier” is a term of art that is used in this specification to contrast embodiments of the optical transmission system disclosed herein with an optical OFDM transmission system. This term should not be interpreted to imply that embodiments of the invention are not compatible with the WDM technology. One of ordinary skill in the art will appreciate that certain signal processing techniques disclosed herein may be applied to each of different carrier frequencies (wavelengths) of a WDM multiplex.
As used herein, the term “synchronous” refers to temporal alignment of two data or symbol blocks, two symbols, and/or the time slots corresponding to them. For example, two symbol blocks are considered to be synchronous if their leading edges arrive at a specified location (e.g., an input port or an output port) substantially simultaneously, i.e., the difference between the times of arrival is smaller than a designated relatively small tolerance. Similarly, two symbols are considered to be synchronous if their leading edges arrive at a specified location substantially simultaneously, i.e., the difference between the times of arrival is smaller than a designated relatively small tolerance.
Although various embodiments of the invention have been described in reference to cyclic prefixes, the invention is not so limited. Based on the provided description, one skilled in the art will be able to similarly practice the invention with cyclic suffixes instead of or in addition to cyclic prefixes. As used herein, the term “guard interval” should be interpreted as a general term that covers both cyclic prefixes and cyclic suffixes.
In various embodiments, each data frame has at least one payload-symbol block (DSi, FIG. 2B) that is concatenated with another block without a guard interval between them. For example, if data frames use cyclic prefixes as guard intervals, then one of such concatenation points is located at the boundary between the last pilot-symbol block of a data frame and the first payload-symbol block of the same data frame, e.g., between pilot-symbol block PSc2 and payload-symbol block DS1 in FIG. 2B. Alternatively, if data frames use cyclic suffixes as guard intervals, then one of such concatenation points is located at the boundary between the last payload-symbol block of a data frame and the first pilot-symbol block of the next data frame, e.g., between payload-symbol block DSn and the adjacent pilot-symbol block PSs in FIG. 2B.
The present invention may be implemented as circuit-based processes, including possible implementation on a single integrated circuit.
Unless explicitly stated otherwise, each numerical value and range should be interpreted as being approximate as if the word “about” or “approximately” preceded the value of the value or range.
It will be further understood that various changes in the details, materials, and arrangements of the parts which have been described and illustrated in order to explain the nature of this invention may be made by those skilled in the art without departing from the scope of the invention as expressed in the following claims.
The use of figure numbers and/or figure reference labels in the claims is intended to identify one or more possible embodiments of the claimed subject matter in order to facilitate the interpretation of the claims. Such use is not to be construed as necessarily limiting the scope of those claims to the embodiments shown in the corresponding figures.
Although the elements in the following method claims, if any, are recited in a particular sequence with corresponding labeling, unless the claim recitations otherwise imply a particular sequence for implementing some or all of those elements, those elements are not necessarily intended to be limited to being implemented in that particular sequence.
Reference herein to “one embodiment” or “an embodiment” means that a particular feature, structure, or characteristic described in connection with the embodiment can be included in at least one embodiment of the invention. The appearances of the phrase “in one embodiment” in various places in the specification are not necessarily all referring to the same embodiment, nor are separate or alternative embodiments necessarily mutually exclusive of other embodiments. The same applies to the term “implementation.”
Also for purposes of this description, the terms “couple,” “coupling,” “coupled,” “connect,” “connecting,” or “connected” refer to any manner known in the art or later developed in which energy is allowed to be transferred between two or more elements, and the interposition of one or more additional elements is contemplated, although not required. Conversely, the terms “directly coupled,” “directly connected,” etc., imply the absence of such additional elements.
The embodiments covered by the claims in this application are limited to embodiments that (1) are enabled by this specification and (2) correspond to statutory subject matter. Non-enabled embodiments and embodiments that correspond to non-statutory subject matter are explicitly disclaimed even if they formally fall within the scope of the claims.
The description and drawings merely illustrate the principles of the invention. It will thus be appreciated that those of ordinary skill in the art will be able to devise various arrangements that, although not explicitly described or shown herein, embody the principles of the invention and are included within its spirit and scope. Furthermore, all examples recited herein are principally intended expressly to be only for pedagogical purposes to aid the reader in understanding the principles of the invention and the concepts contributed by the inventor(s) to furthering the art, and are to be construed as being without limitation to such specifically recited examples and conditions. Moreover, all statements herein reciting principles, aspects, and embodiments of the invention, as well as specific examples thereof, are intended to encompass equivalents thereof.
The functions of the various elements shown in the figures, including any functional blocks labeled as “processors,” may be provided through the use of dedicated hardware as well as hardware capable of executing software in association with appropriate software. When provided by a processor, the functions may be provided by a single dedicated processor, by a single shared processor, or by a plurality of individual processors, some of which may be shared. Moreover, explicit use of the term “processor” or “controller” should not be construed to refer exclusively to hardware capable of executing software, and may implicitly include, without limitation, digital signal processor (DSP) hardware, application specific integrated circuit (ASIC), field programmable gate array (FPGA), read only memory (ROM) for storing software, random access memory (RAM), and non volatile storage. Other hardware, conventional and/or custom, may also be included. Similarly, any switches shown in the figures are conceptual only. Their function may be carried out through the operation of program logic, through dedicated logic, through the interaction of program control and dedicated logic, or even manually, the particular technique being selectable by the implementer as more specifically understood from the context.
It should be appreciated by those of ordinary skill in the art that any block diagrams herein represent conceptual views of illustrative circuitry embodying the principles of the invention. Similarly, it will be appreciated that any flow charts, flow diagrams, state transition diagrams, pseudo code, and the like represent various processes which may be substantially represented in computer readable medium and so executed by a computer or processor, whether or not such computer or processor is explicitly shown.

## Claims

1. An optical receiver, comprising:
an optical-to-electrical converter adapted to:
mix a modulated optical signal with an optical reference signal to generate a plurality of mixed optical signals; and
convert the plurality of mixed optical signals into a corresponding plurality of digital electrical signals, wherein:
the plurality of digital electrical signals comprises a first in-phase digital signal and a first quadrature-phase digital signal that have a first set of signal samples corresponding to a data frame;
the data frame comprises:
a first plurality of pilot-symbol blocks, wherein each pilot-symbol block comprises a respective plurality of pilot symbols and a respective guard interval; and
a first set of one or more payload-symbol blocks, wherein each payload-symbol block comprises a respective plurality of constellation symbols from a first sequence of constellation symbols; and

at least one payload-symbol block and another block are concatenated without a guard interval between them, wherein the other block is either a pilot-symbol block or a payload-symbol block; and


a digital signal processor adapted to process the first set of signal samples to recover data encoded in the first sequence of constellation symbols.

2. The optical receiver of claim 1, wherein the first set of one or more payload-symbol blocks comprises a plurality of payload-symbol blocks, wherein each pair of consecutive payload-symbol blocks in the first set is concatenated without a guard interval between them.

3. The optical receiver of claim 1, wherein:
the first plurality of pilot-symbol blocks comprises a frame-synchronization block having a first string of N pilot symbols concatenated with a second string of N pilot symbols, said second string being a copy of the first string, where N is a positive integer; and
the digital signal processor is further adapted to:
calculate an autocorrelation function for the first set of signal samples; and
find a maximum of the autocorrelation function to determine a temporal position of the data frame.

4. The optical receiver of claim 3, wherein the digital signal processor is further adapted to:
calculate a phase of the autocorrelation function at the maximum; and
calculate a frequency offset between a carrier frequency of the modulated optical signal and a frequency of the optical reference signal based on the calculated phase; and
apply frequency-offset correction to the first set of signal samples to produce a first modified set of signal samples, wherein each signal sample of the first modified set has a phase shift with respect to a corresponding signal sample of the first set of signal samples that is proportional to the calculated frequency offset and relative time of the signal sample in the first set of signal samples.

5. The optical receiver of claim 1, wherein:
the first plurality of pilot-symbol blocks comprises:
a first channel-estimation block having a first specified polyphase sequence of symbols; and
a second channel-estimation block having a second specified polyphase sequence of symbols different from the first specified polyphase sequence of symbols; and

the digital signal processor is further adapted to determine a channel-response function H based on signal samples of the first set of signal samples that correspond to said first and second polyphase sequences.

6. The optical receiver of claim 5, wherein:
the plurality of digital electrical signals further comprises a second in-phase digital signal and a second quadrature-phase digital signal that have a second set of signal samples corresponding to the data frame;
the data frame further comprises:
a second plurality of pilot-symbol blocks, wherein each pilot-symbol block comprises a respective plurality of pilot symbols and a respective guard interval; and
a second set of one or more payload-symbol blocks, wherein:
each payload-symbol block of the second set comprises a respective plurality of constellation symbols from a second sequence of constellation symbols;
a first polarization of the modulated optical is modulated with the symbols of the first plurality of pilot-symbol blocks and the first set of one or more payload-symbol blocks; and
a second polarization of the modulated optical is modulated with the symbols of the second plurality of pilot-symbol blocks and the second set of one or more payload-symbol blocks.

7. The optical receiver of claim 6, wherein:
the first plurality of pilot-symbol blocks is synchronous with the second plurality of pilot-symbol blocks; and
the first set of one or more payload-symbol blocks is synchronous with the second set of one or more payload-symbol blocks.

8. The optical receiver of claim 6, wherein:
the second plurality of pilot-symbol blocks comprises:
a third channel-estimation block having the second specified polyphase sequence; and
a fourth channel-estimation block having the first specified polyphase sequence;

the first channel-estimation block is synchronous with the third channel-estimation block;
the second channel-estimation block is synchronous with the fourth channel-estimation block; and
the digital signal processor is further adapted to determine the channel-response function H based on signal samples of both the first and second sets of signal samples that correspond to the first and second specified polyphase sequences.

9. The optical receiver of claim 6, wherein the digital signal processor is further adapted to apply an inverse channel-response function H−1 to produce a first modified set of signal samples and a second modified set of signal samples, wherein:
the first modified set has contributions from both the first and second sets of signal samples; and
the second modified set has contributions from both the first and second sets of signal samples.

10. The optical receiver of claim 1, wherein the digital signal processor is further adapted to:
apply frequency-offset correction to the first set of signal samples to produce a first sequence of frequency-offset corrected signal samples corresponding to the first plurality of payload-symbol blocks;
select a first block of frequency-offset corrected signal samples in the first sequence;
apply channel-response equalization to the first block of frequency-offset-corrected signal samples to produce a first block of channel-response-equalized signal samples;
remove a sub-block of signal samples from a start portion of the first block of channel-response-equalized signal samples and another sub-block of signal samples from an end portion of the first block of channel-response-equalized signal samples to produce a first truncated block of signal samples;
select a second block of frequency-offset corrected signal samples in the first sequence, wherein the second block overlaps with the first block;
apply channel-response equalization to the second block of frequency-offset-corrected signal samples to produce a second block of channel-response-equalized signal samples;
remove a sub-block of signal samples from a start portion of the second block of channel-response-equalized signal samples and another sub-block of signal samples from an end portion of the second block of channel-response-equalized signal samples to produce a second truncated block of signal samples; and
concatenate the first truncated block and the second truncated block.

11. The optical receiver of claim 10, wherein, to apply channel-response equalization, the digital signal processor:
applies a Fourier transform to a selected block of signal samples to produce a corresponding block of frequency components;
applies an inverse channel-response function H−1 to the block of frequency components to produce a block of channel-response-equalized frequency components; and
applies an inverse Fourier transform to the block of channel-response-equalized frequency components to produce a corresponding block of a channel-response-equalized signal samples.

12. The optical receiver of claim 10, wherein:
each of the first and second blocks of frequency-offset corrected signal samples has 2N signal samples, where N is a positive integer;
each removed sub-block has L/2 signal samples, where L is a positive even integer;
the cyclic prefix in each of the pilot-symbol blocks in the data frame has L pilot symbols; and
each payload-symbol block in the data frame has 2N−L constellation symbols.

13. An optical communication method, comprising:
mixing a modulated optical signal with an optical reference signal to generate a plurality of mixed optical signals;
converting the plurality of mixed optical signals into a corresponding plurality of digital electrical signals, wherein:
the plurality of digital electrical signals comprises a first in-phase digital signal and a first quadrature-phase digital signal that have a first set of signal samples corresponding to a data frame;
the data frame comprises:
a first plurality of pilot-symbol blocks, wherein each pilot-symbol block comprises a respective plurality of pilot symbols and a respective guard interval; and
a first set of one or more payload-symbol blocks, wherein each payload-symbol block comprises a respective plurality of constellation symbols from a first sequence of constellation symbols; and

at least one payload-symbol block and another block are concatenated without a guard interval between them, wherein the other block is either a pilot-symbol block or a payload-symbol block; and

processing the first set of signal samples to recover data encoded in the first sequence.

14. The method of claim 13, wherein:
the first plurality of pilot-symbol blocks comprises a frame-synchronization block having a first string of N pilot symbols concatenated with a second string of N pilot symbols, said second string being a copy of the first string, where N is a positive integer; and
the processing comprises:
calculating an autocorrelation function for the first set of signal samples; and
finding a maximum of the autocorrelation function to determine a temporal position of the data frame.

15. The method of claim 13, wherein:
the first plurality of pilot-symbol blocks comprises:
a first channel-estimation block having a first polyphase sequence of symbols; and
a second channel-estimation block having a second polyphase sequence of symbols different from the first polyphase sequence of symbols; and

the processing comprises determining a channel-response function H based on signal samples of the first set of signal samples that correspond to the first and second polyphase sequences, wherein:
the plurality of digital electrical signals further comprises a second in-phase digital signal and a second quadrature-phase digital signal that have a second set of signal samples corresponding to the data frame;
the data frame further comprises:
a second plurality of pilot-symbol blocks, wherein each pilot-symbol block comprises a respective plurality of pilot symbols and a respective guard interval; and
a second set of one or more payload-symbol blocks, wherein:
each payload-symbol block of the second set comprises a respective plurality of constellation symbols from a second sequence of constellation symbols;
a first polarization of the modulated optical is modulated with the symbols of the first plurality of pilot-symbol blocks and the first set of one or more payload-symbol blocks; and
a second polarization of the modulated optical is modulated with the symbols of the second plurality of pilot-symbol blocks and the second set of one or more payload-symbol blocks.

16. The method of claim 15, wherein:
the second plurality of pilot-symbol blocks comprises:
a third channel-estimation block having the second polyphase sequence; and
a fourth channel-estimation block having the first polyphase sequence;

the first channel-estimation block is synchronous with the third channel-estimation block;
the second channel-estimation block is synchronous with the fourth channel-estimation block; and
the channel-response function H is determined based on signal samples of both the first and second sets of signal samples that correspond to the first and second polyphase sequences.

17. The method of claim 15, wherein the processing comprises applying an inverse channel-response function H−1 to produce a first modified set of signal samples and a second modified set of signal samples, wherein:
the first modified set has contributions from both the first and second sets of signal samples; and
the second modified set has contributions from both the first and second sets of signal samples.

18. The method of claim 13, wherein the processing comprises:
applying frequency-offset correction to the first set of signal samples to produce a first sequence of frequency-offset corrected signal samples corresponding to the first plurality of payload-symbol blocks;
selecting a first block of frequency-offset corrected signal samples in the first sequence;
applying channel-response equalization to the first block of frequency-offset-corrected signal samples to produce a first block of channel-response-equalized signal samples;
removing a sub-block of signal samples from a start portion of the first block of channel-response-equalized signal samples and another sub-block of signal samples from an end portion of the first block of channel-response-equalized signal samples to produce a first truncated block of signal samples;
selecting a second block of frequency-offset corrected signal samples in the first sequence, wherein the second block overlaps with the first block;
applying channel-response equalization to the second block of frequency-offset-corrected signal samples to produce a second block of channel-response-equalized signal samples;
removing a sub-block of signal samples from a start portion of the second block of channel-response-equalized signal samples and another sub-block of signal samples from an end portion of the second block of channel-response-equalized signal samples to produce a second truncated block of signal samples; and
concatenating the first truncated block and the second truncated block.

19. The method of claim 18, wherein the channel-response equalization comprises:
applying a Fourier transform to a selected block of signal samples to produce a corresponding block of frequency components;
applying an inverse channel-response function H−1 to the block of frequency components to produce a block of channel-response-equalized frequency components; and
applying an inverse Fourier transform to the block of channel-response-equalized frequency components to produce a corresponding block of a channel-response-equalized signal samples.

20. The method of claim 18, wherein:
each of the first and second blocks of frequency-offset corrected signal samples has 2N signal samples, where N is a positive integer;
each removed sub-block has L/2 signal samples, where L is a positive even integer;
the guard interval in each of the pilot-symbol blocks in the data frame has L pilot symbols; and
each payload-symbol block in the data frame has 2N−L constellation symbols.