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VOLUME la 

TECHNICAL 

PROPOSAL 

5 JULY 1968 

BETA SYSTEM 

MODEL HG-469B 

Neon 

700 ROYAL OAKS DRIVE 

MONROVIA, CALIFORNIA 

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, SECRETED 

c BIF-005-112-68 
BIF-005-112-68 

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Volume II 

CONTENTS 

Page 

1. BETA SYSTEM REQUIREMENTS 1-1 

1.1 Introduction 1-1 

1.2 Performance 1-1 

1.3 Acceptance and Qualification Testing 1-6 

2. THEORY AND BACKGROUND 2-1 

2.1 Introduction 2-1 

2.2 Basic Requirements 2-1 

2.3 Terrain Light Patterns 2-2 

2.4 Generation of the Photosignal 2-5 

2.5 Apertures and Spectral Properties 2-8 

2.6 Extracting Velocity Information 2-21 

2.7 Error 2-24 

2.8 Comparison of Methods 2-46 

3. DESIGN APPROACH 3-1 

3.1 General Description 3-1 

3.2 Detector 3-2 

3.3 Electronics Package 3-5 

3.4 Design Data 3-7 

3.5 Equipment Characteristics 3-21 

3.6 Specification Compliance 3-27 

3.7 The Problem, Risks, and Solutions 3-30 

3.8 A/D Converter 3-33 

3.9 Testing 3-33 

ii 
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Page 

4. MAINTAINABILITY AND RELIABILITY 471 

4.1 Maintainability 4-1 

4.2 Reliability 4-3 

APPENDIX 1. BETA SIGNAL SLOPE LIMITER 

APPENDIX 2. HERRINGBONE IMAGE RATE SENSOR 

• 

iii 

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Section 1 

BETA SYSTEM REQUIREMENTS 

1.1 INTRODUCTION 

The Hycon Company is pleased to submit this proposal to manufacture and 

deliver four engineering models, four preproduction units, and twelve production units of the Beta System, Model HG-469B, including those retained 

for testing. The general theory of image motion detection systems is 

described in Section 2 and the specific Hycon system is detailed in Section 

3. The Hycon Beta System is the outgrowth of an extensive in-house 

research program which resulted in the HG-453 production model single axis 

sensor, shown in Figure 1-1. Section 4 is devoted to maintainability, 

safety, and reliability. 

1.2 PERFORMANCE 

Principle performance characteristics of the Beta System, as detailed in 

Specification EC-701A, Rev. 6, form the basis for the Hycon Beta System, Model 

HG-469B, Figure 1-2. 

1.2.1 Functional Characteristics. The HG-469B will sense image velocity 

at the center of the format of the scene image and resolve it into two 

vector components along perpendicular axes X and Y. The HG-469B will provide 

the following output signals: 

a. Image Velocity 

b. Lock-on Signal 

c. Operational Readiness Monitoring 

d. Diagnostic Monitoring 

e. Subthreshold image plane irradiance indication 

1-1 

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Page__ 

Figure 1-1. HG-453 Production Model Sensor 

1-2 

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ELECTRONICS PACKAGE 

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ELECTRONICS PACKAGE 

Figure 1-2. 

Beta System, Model HG-469B 

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1.2.2 Primary Performance Characteristics 

1.2.2.1 Image Velocity Output Signals. The output image velocity 

signals of the Beta System will be proportioned to the components of the 

image velocity vector resolved along the reference, perpendicular axes. 

The reference axes will be indexed coincident to the Beta System base. 

The image velocity signal for each of the two axes will be of analog 

form with a maximum of +5 VDC. A separate sense or sign signal shall 

be provided for each axis. The nominal gradient will be 16.67 volts/ 

inch/second. 

a. Dynamic Range. The Beta System will provide single-valued voltages 

representative of true image velocities from 0 to 0.3 inch/second (600 micro 

rad/sec) in the primary image plane. Image rotational velocity in the image 

Mane will not be sensed. 

b. Linearity. The relationship between true image velocity and indicated image velocity will be of the form V-True = K x V-Ind for true image 

velocity between 0.0225 inch/sec (45 micro rad/sec) and 0.3 inch/sec (600 

micro rad/sec) where: 

V-True = true image velocity and 

V-Ind = indicated image velocity 

The three sigma limits of K including noise are 0.738 and 1.55. Signalto-noise ratio for the image velocity range stated will be greater than two. 

c. Saturation. The Beta System will not saturate for true image 

velocities of less than 0.3 inch/sec (600 micro rad/sec). When saturated, the 

Beta System will provide a signal indicating this condition for each axis. The minimum for this condition will be a polarity signal corresponding to the velocity 

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polarity for velocities up to 0.5 inch/sec. The Beta System will be capable 

of recovering within 0.5 seconds of receiving an input within its dynamic 

range from saturation periods up to two seconds. 

d. Frequency Response. The dynamic response of the Beta System over 

the frequency range from 0 to 60 rad/sec will resemble that of a device with 

a single lag located between 6 rad/sec and 60 rad/sec. The desired location 

of the lag is at 7 rad/sec. In addition a notch filter will be located at 

188.4 rad/sec. 

e. Null Accuracy. When the true image velocity is equal to or less 

than 0.0225 inch/sec (45 micro rad/sec), the indicated velocity, as provided 

by the Beta System will be in error by no more than 0.010 inch/sec (20 micro 

rad/sec). 

1.2.2.2 ' Lock-on Signal. The Beta System will provide a signal for 

each axis indicating the image lock-on status. The output will be 3.5 to 5.0 

vdc when the image is in the locked-on condition and 0.5 +.5 vdc when not locked 

or saturated. 

1.2.2.3 Operational Readiness Signal. The Beta. System will provide 

an output signal to denote operational readiness. 

a. The signal magnitude will be a 3.5 to 5.0 vdc when it is operationally 

ready. 

b. The signal magnitude shall be 0.5 to +0.5 vdc when it is not operationally ready. 

1.2.2.4 Diagnostic Monitoring Signals. Necessary diagnostic monitoring 

signals will be provided for telemetry monitoring as required. The test points 

will provide continuous output signals within the range of 0 to 5 vdc. 

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1.2.2.5 Subthreshold Irradiance Signal. The Beta will provide an 

output signal to indicate when the average input irradiance in the primary 

image plane is insufficient for proper Beta performance. 

a. The signal magnitude will be 3.5 to 5.0 vdc when sufficient light 

exists. 

b. The signal magnitude will be 0.5 +0.5 vdc when insufficient 

light exists. 

1.3 ACCEPTANCE AND QUALIFICATION TESTING 

1.3.1 An extensive design evaluation and test program will be performed 

according to applicable portions of the Integrated Test Plan which has been 

prepared in accordance with.MOM-T-101-1. Hycon's Integrated Test, Plan is 

presented in the Mangement Volume (Volume I), Appendix A2. 
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Section 2 

THEORY AND BACKGROUND 

2.1 INTRODUCTION 

This section describes the basis of operation of typical image velocity 

sensors in terms which lead to a general understanding of methods. This allows 

comparison of known methods and enlightens the search for improved types. The 

operation of such systems is best treated by mathematical theory which makes use 

of basic statistical principles and the Fourier Transform relationships between 

power spectra and correlation functions. This section presents the elements of 

the theory in the form of results, the detailed origin of which may be found in 

• 

cited reports and references, in a sequence which leads to the needed generalized 

understanding of image velocity measurement techniques. 

2.2 BASIC REQUIREMENTS 

The Beta System is required to provide image velocity information which may 

be used, either in open-loop or closed-loop systems, to minimize resultant image 

plane relative motion, that is, to achieve image motion compensation (1MC). In 

many cases, the image plane shear during the exposure interval is small to begin 

with, so that high percentage accuracy in Beta System operation is not a primary 

aim. However, very little drift or null error can be tolerated. 

In addition to null stability and small null error, speed of response is very 

important in permitting adequate corrections for image velocity changes resulting 

- from platform or vehicle attitude perturbations or altitude change with flight. 

Since platform motion may be relatively great prior to exposure, the Beta System 

must not be subject to saturation from large image velocities, or must recover 

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very rapidly from such saturation. Adequate velocity range, with good linearity, 

is necessary in closed-loop operation for stable, rapid covergence to null velocity. Saturation or inadequate gain (scale) uniformity will generally lead to 

. instability in the form of blocking (nonlinear) oscillations or sustained hunting. 

The use of the image velocity sensor in a servo loop imposes additional requirements on its transfer phase and gain, and, further demands at least nominally 

linear operation. 

The fundamental notion is that image velocity can be corrected if it can be 

measured. In reality, different parts of the image may have different velocities 

because of altitude variations within the field of view, or because of slant 

range distribution resulting from large obliquity angles. Where image velocity 

is such a velocity distribution, the image velocity sensor can be expected to 

provide a weighted average image velocity. Determination of the optimum field 

of view for the Beta system involves consideration of terrain and altitude 

statistics. 

Because of the one-time character of most photographic missions, reliability 

outranks detailed performance as a requirement. In seeking reliability, one not 

only tries to stay with well-known, highly developed techniques, but also seeks 

simplicity in operating principle and make-up. Fortunately, great improvements 

have been made in reliability of electronic circuits, largely through development 

of very rugged, compact semiconductor devices. With such components, necessary 

circuit functions can be implemented with very small size, weight, and power, 

limitations. 

2.3 TERRAIN LIGHT PATTERNS 

The basic information input to the sensor is the pattern of terrain light 

2-2 

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and shade within the instantaneous field of view. prightness or luminance (L) 

of the terrain may be expressed in footlamberts or equivalent units. We will 

assume the pattern is spatially random, average luminance is Lo, and fluctuations 

2 

of brightness about the average is normally distributed with variance <1 . The 

quantity 01 represents the amount of useful information in the terrain pattern, 

and may be referred to as terrain power. Figure 2-1 presents a one-dimensional 

terrain luminance distribution. 

It will be important in our considerations to know, or to assume, the manner 

in which this terrain power is distributed as a function of spatial wave numbers. 

The terrain-power spectrum PT is the (two-dimensional) function of (two) spatial 

wave numbers which represents the density (over +00 to -00 in each of the wave 

number variables), excluding a small region at the origin (due to L0), is the 

total terrain power CI . A one-dimensional power spectrum is sketched in 

figure 2-2. 

We are not in a position to characterize PT very well; even if one had a 

typical terrain power spectrum, it is certain that it would be found atypical 

for many terrain scenes. This is a philosophical problem rather than statistical 

ignorance. The performance of a Beta system depends, in some cases critically, 

upon the amount of terrain power falling within prescribed regions of the terrain 

power spectrum, and performance estimates are therefore sensitive to the functional • 

form assumed. If photographs of typical terrain specimens were available for 

analysis, a few samples of terrain spectra could be computed; however, neither 

such samples nor hypotheses can bridge the gap between the a'posteriori results 

of experiments and the a'priori estimation of subsequent trials. 

For purposes of discussion, assume that the terrain power spectrum has the 

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—72---)X kel,%9414.14: 

X-6") 

/4N4/ 

.2),sprhard774V 

7t.5,774,0i/ 

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Figure 2-1. Hypothetical Luminance Pattern for a One-dimensional. 

Terrain. (Terrain Power . CFI! ). 

0/21E 72) 4 • 

.1144 401" 

Figure 2-2. One-dimensional Terrain Power Spectrum. 

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form 

P P ---14 2 )

n 

T T 0 U2"4- 1.1 

ti2 + 

(2.1) 

where n >1 in order that the total power be finite. The delta function 

represents the average component. When the integrations are performed, we 

obtain 

( n - I ) 

• 7T14 

It is usual to characterize the variations in luminance of well defined 

targets, such as bar charts used for resolution testing, by the contrast ratio, 

defined as the ratio of L max to L min (L max being typical of a bright element 

and L min the dark element). With random patterns having normal or near-normal 

distributions of brightness, this characterization must be reinterpreted in a 

very subjective manner. To make contact between CI, representing the normal 

distribution of luminance in the terrain, and the contrast ratio, we may choose 

for L max that value which is exceeded only rarely, and a similar value on the 

lower end for L min. If we take these bounds at +eCronrarelyn is 2.3% of the 

time. For this condition: 

Contrast Ratio L04.2q_ 

Lo 2 L0 L0 

For example, a contrast ratio Of 1.2:1 would correspond to CTL = • L0/20 

2.4 GENERATION OF THE PHOTOSIGNAL 

The Beta system receives the image of the terrain as formed by an optical 

system. Each element of the terrainbf L in footlamberts is imaged 

as a corresponding element of illuminance I in footcandles, where 

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(transmission factors) (2.4) 

f being the f-number (relative aperture) of the optical system. The transmission factors represent loss due to atmospheric scattering and optical 

reflectance, and due to transmission losses. 

The field of view of the Beta system is defined by an aperture, which may 

have any shape and may be shaded or apodized to improve performance. All of 

the image light transmitted by the aperture is collected and/or falls upon a 

photosensor of sensitivity S (amperes per watt). The resulting photocurrent or 

signal is thus proportional to the integral of the product of image illuminance 

and aperture transmission, integration being taken over the entire aperture. As 

the image moves with respect to the aperture, the weighting function, represented 

by the aperture transmittance, scans across the image, and the resulting photo 

signal fluctuates accordingly. The properties of this fluctuating signal depend 

in a concisely definable manner upon the power spectra of the terrain and upon 

the aperture. 

The integral of the product of a weighting function and a second function in 

which the weighting function shifts along the axis of variables while the second 

function remains fixed in position is well known in many phases of mathematics, 

where the operation is called convolution or folding. Furthermore in transform 

theory, convolution is shown to be equivalent to the product of the transforms 

of the two functions. These relations become particularly direct and useful to 

us in the domain of Fourier transforms. As a consequence of these known principles, it can be shown that the power electrum of the photosignal is proportional 

to the product of the power spectra of th;.! aperture and of the terrain, with 

suitable scaling for velocity (in going from wave number to frequency), as shown 

in figure 2-3. 

In estimating performance of an imago velocity system, the error in velocity 

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Figure 2-3. Generation of Photosignal E(s). (Passing terrain image 

illumination through aperture A(x). The corresponding 

multiplication of power spectra to obtain Poi; is illustrated 

at the bottom of the Noire.) 

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measurement is derived from the power spectrum of the photosignal. For this 

reason we are concerned with power spectra rather than amplitude spectra. The 

convolution theorem of Fourier transforms is not• only an important tool, but is 

sufficient for our analysis. 

Photosensors differ, from the point of view of our analysis, primarily in 

sensitivity and noise properties. Photomultiplier tubes have relatively low 

sensitivities, but the high gains achieved by electron multiplication permit 

operation of subsequent stages at low impedance. This virtually eliminates the 

source of thermal noise but shot noise remains, increased somewhat by dynode 

action. Photodiodes show higher sensitivities, but include thermal noise and 

shot noise in consideratioh. As light level decreases, the choice of photosensor 

may shift from photodiode to photomultiplier because of thermal noise advantage. 

2.5 APERTURES AND SPECTRAL PROPERTIES 

2.5.1 Spectral Properties 

Since the spectrum- of the aperture combines with that of the terrain (image) 

to determine performance, we now consider the properties of several types of 

apertures. The distinctions between various diverse categories of image velocity 

instruments can be best understood in terms of the spectral properties of the 

apertures and their interaction with the image velocity and spectrum. 

Each static aperture may be represented by a transmission which is a function 

of two spatial dimensions, x in the forward direction and y in the transverse 

direction. Let L and p. be the corresponding conjugate wave numbers. Then 

the amplitude spectrum of the aperture is a function of both t, and JL , given 

by the Fourier 'Transform 

oc' +co 2 TrIf x vi(v,11).--, if A (x,y)re dxcly. (2.5) 

6-EGRET-tia- 2-8 hp_ /17 of 
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in which A(x,y) is the aperture transmission andAWis the aperture amplitude 

spectrum. The aperture power spectrum is equal to the square of the absolute 

value of the amplitude spectrum: 

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- PA ( ) = 2)01 ) 

As an example, consider a rectangular aperture of forward length 

(2.6) 

and transverse width W . The power spectrum of such a uniform rectangular 

aperture has the form 

f) 

sin -TrlAtv sin7rNLE) 

• 

(1-q / )7= • A • 7T/I -apt ./ • (2.7) 

which is illustrated in figure 2-4. Since the photosignal resulting from the 

use of this aperture is a convolution of the aperture with the image, we know 

by the convolution theorem of Fourier transforms that the amplitude spectrum of 

the photosignal is equal to the product of the amplitude spectra of the aperture 

and the image (or terrain, scaled for the optical demagnification). 

But we don't know the amplitude spectrum of- the image; we know (even in 

principle) only the power spectrum. And we don't want the amplitude spectrum 

of the photosignal; we want only the power spectrum. 

Let us consider the formation of the photosignal in more detail. The signal 

EMis given by the integral 

40::. 4- co 

ff d X -- f ,y ) I x,y dxdv CL ' 

This integral may be described as the integral of the product of two quantities. 

Going now to Fourier transforms, we may matte an. entirely equivalent statemeht: 

the transform t: 6) is equal to the values lying on the f axis of the convolution 

2-9 

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Figure 2-dt. One-dimensional Apekure A:aplitude and Power Spectra. 

2-10 

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of 4(11,11 f) and 1O1,11; f J : 

E(0-z-- v,μ;f) 31 ( f v,= o . (2.9) 

where the asterisk is a conventional symbol for convolution. Because time 

enters into equation (2.8) an well as the spatial variables x and y, we must 

operate in a:three-dimensional transform space involving f, V, and 11. The 

convolution is over this entire space. But the manipulations are made simple 

by the fact that I(x,y) is not a function of time and, therefore, its transform 

1. (z4L.,f)exists only in the f=0 plane. FUrther, since the aperture moves linearly 

".1 

with velocity Va, the transformA(, ;f)exists only on a plane f=VaLI. These 

restrictions result in the full convolution process reducing to a simple prof-, 

jection of I 01)0 

weighting f1(11,4). Figure 2-5 illustrates this process of derivation. 

The fact remains that, in projecting I( Up.) by means oftkWonto the f 

axis to form E.(f), many different elements of' NO contribute to each 

element of EN. When we square [nto create PE we get cross products in 

addition to sums of squares. But it is usual to assume that the components of 

power spectra are not mutually correlated, and on this basis we may expect the 

cross terms to vanish in the average. Therefore, the square of the projected 

sums of products. is equal to the sum of the projected product of squares, or 

r(f) = PA (1;11 f) * Pr( /1i; f) , (2.10) 

Hence, we need concern ourselves only with the power spectra and with their 

multiplication and reflection in the plane of slopcVa, as determined by the 

motion. 

For a simple fixed aperture showing no periodic structure, the aperture trans-

- form lies about the origin on the locus plane f -41,119 . The phOtdsIgnal then 

onto the f axis by reflection in the plane f:=Vciiii with 

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Figure 2-5. One-dimensional CaSe. (Projection of I (0 into E(f) by means 

of aperture transform (L),f , the locus of which is determined by 

the aperture velocity % (relative to the image). Because. j CO 

exists only along lJ axis, and 7k(v,f)exists only along f =Vctv , 

convolution becomes a simple projection. In more complex situations, 

this simplification does not hold.) 

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has a power spectrum which is concentrated about f=0 and has width proportional 

to Va 

Now consider an aperture which consists of a square-wave grid of density within 

the normal outline of the aperture. The aperture transform consists of replications of the transform of the outline spaced along the locus plane f=t/av at 

intervals equal to Vc3 , the grid wave number. The amplitude of the successive 

images are equal to the auplitudes'of the harmonics of the wave structure within 

the aperture. If a sinusoidal grid is used there will be three patches of 

density on the locus plane: one at the origin, of amplitude 1/2 representing the 

average transmission; and one at (4-2),j)0), of amplitude 1/4 corresponding to the 

two cisoidal components. 

As velocity Vachanges, the locus plane tilts accordingly (with tilt components 

Vax and V

ay 

 ) and the narrow band projections move up and down the frequency axis. 

If the aperture is not extended, as in the fixed-grid case, but is highly 

localized as is a pinhole or slit, the aperture transform becomes very broad, 

and its projection yields an EP which is correspondingly broadband. 

To this point, we have mentioned only fixed apertures in which velocity Va 

appears only because of the motion of the aperture across the image. Such aper- 

. • 

tures are evidently worthless under null-velocity conditions and they could not 

be Used for servo image stabilization. The treatment becomes more interesting 

when we allow the aperture to become dynamic. Suppose that the grid moves 

continuously across the aperture outline at a high velocity Vg. The transform 

of this aperture exist on planes of slope Vacentered along a line of slope Vg. 

A / The projections of the transform density /A pili)upon the f=0 plane are the same /I 

as for Vg=0, the fixed grid case, but the frequencies of the harmonics are moved 

out harmonic multiples of \/9 Vs. beyond their former positions. Now the spectrum 

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E (f) does not collapse to zero frequency as Va goes to zero. As the locus 

planes f = Va v tilt with velocity, the narrowband peaks of E (I') move about 

a new origin f = V

gvg 

 (for the sinusoidal grid). Hence the failure of operation at null velocity and the ambiguity in sign is entirely avoided by moving 

the grid within the aperture. This is the basis of the phase-rate-measurement. 

2.5.2 Aperture Modulation 

Optimum design of a spatial filter IVS requires careful consideration of 

aperture shape and shading in relation to grid spatial frequency. The choice 

of grid spatial frequency is relatively easy to make, being based on observation 

of measured spectra of typical scenes. Not so obvious, however, is the choice 

of "best" aperture shape and apodization (shading). 

Previous studies have established that a circular aperture is best because 

it results in minimum noise due to transverse image velocity components. This 

section derives the optimum relationship between grid spatial frequency and 

aperture diameter by means of Fourier Transform methods, and establishes the 

attenuation factor for D.C. aperture modulation in terms of grid wave number, 

aperture diameter, and the degree of Gaussian edge diffusion. 

"Aperture modulation" is a cogent consideration because it can be the 

limiting factor in the ability of an IVS to function properly under conditions 

of high scattered light (haze) and/or low scene contrast. Aperture modulation 

manifests itself in the following forms: 

a) "scalloping" or periodic low frequency noise superimposed on the output 

voltage of the IVS 

apparent "drop-outs" 

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c) in the extreme case, complete inability to discern scene motion in the 

presence of overwhelming "haze" light. 

Fortunately, aperture modulation can be reduced a million-fold by employing 

a moderate degree of aperture edge gradient diffusion. 

In the Hycon Beta System, aperture shading can be effected either at the 

input field lens or in proximity to each channel's photomultiplier tube. The 

use of a single common shading mask at the field lens has the advantage of 

simultaneously compensating both the X and Y channels, while protecting the 

field lens from dirt and scratches. Individual X and Y aperture masks; in 

the form of a small hole punched in a thin metal plate and situated between 

the disc and each photomultiplier tube,.would. be an alternate solution. 

2.5.3 Aperture Response to Mean Illumination 

The response of the IVS to background or mean illumination can be understood 

and computed in terms of the aperture amplitude spectrum and the terrain amplitude spectrum. Figure 2-6 illustrates the manner in which the terrain amplitude 

spectrum may be projected, by reflection in the aperture amplitude spectrum, 

upon the temporal frequency axis to yield the amplitude spectrum of the photodetector signal. While the amplitude spectra of the terrain and the output 

signal are complex, for purposes of discussion and illustration we show only 

the moduli. in Figure 2-6. 

The IVS may be considered to consist of a grid, moving at velocity Vg (with 

respect to a stationary terrain), upon which is superimposed a (circular) 

aperture moving at a much slower velocity V'. The grid transmission is a 

a 

periodic squarewave of spatial frequency li- g. The radius of the circular 

aperture is a. 

2-15 

-SEGRater 
2-6. Illustrating the Derivation of Output Spectrum By Projecting 

the Terrain Illumination Spectrum Onto Frequency Axis by Means 

_.....sfeRata- of the Aperture Spectrum. 

2-16 -g! 

Figure 

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The amplitude spectrum of the combined aperture-grid is found by convolving 

the spectra of the moving aperture with that of the rapidly moving grid. When 

this is done, it is found that the density of the combined aperture-grid amplitude spectra lie wholly along lines of slope Va which intersect the frequency 

axis at multiples of Vg vg = fo, where fo is the "carrier frequency" of the 

IVS system. Because the rectangular grid has only zeroth plus odd harmonies, 

there is one such line of slope. through the origin, other through + f0, - +3f0, 

+5f0, etc. However, a filter is used to eliminate signal components outside 

a narrow region about +f0. This region is shown, grossly exaggerated, in 

Figure 2-6 as "crystal filter bandpass". The slope is also very greatly 

exaggerated in the figure. 

We have said that the aperture amplitude spectrum exists only on the lineS 

of slope Va. Along these lines, the amplitude spectral density is distributed 

in the exact form of the amplitude spectrum of the circular aperture alone 

(without the grid) with the origins shifted by multiples of vg. Thus a weight 

function of peak magnitude na2

/2 occurs about the origin due to the D.C. component of grid transmission, another falls at + vg(on the line of slope Va 

through f = +10 ) due to the fundamental component of the grid, its amplitude 

being Tra2/77 . These features are shown in Figure 2-6 labeled Ao(v) and 

Al ( v ). 

The output signal spectrum E(f) is obtained by taking the value of the terrain amplitude spectrum T(v), multiplying it by Al(v) {for the fundamental 

casell, and replotting the resulting product as E(f) where f = v (Vg Va). 

Al(V) is highly localized to the region close to vg; hence, we expect most 

of the signal to consist of components near fx= vg(Vg + Va). But the terrain 

amplitude spectrum invariably rises rapidly near the origin of v , and it may 

2-17 

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have a large singular peak at the origin due to mean illumination, especially 

if such background illumination is added. Therefore, it is possible for the 

product of the large D.C. amplitude of T(vzo) and the residual ripples of 

A1(v o) to have a resultant magnitude comparable to A1( v g)T( v g) = E( v g) 

if special care is not taken to keep the ripples of A1( o) very small. 

When D.C.-feedthrough occurs, it appears as a photodetector output component at the carrier frequency fo which beats with the desired component at 

fx. When the "feedthrough" is small, the effective frequency of the output 

wave is sinusoidally modulated at the difference frequency; for larger feedthrough, the modulation becomes quite non-linear. The effects can be computed 

readily by analyzing the rotation of the vector resultant of the superposition 

of a fixed and a rotating vector, 

The amount of feedthrough depends upon the magnitudes of A1(v) and T(v). 

Since T(v) is invariably much larger at v= o than elsewhere, we may concern 

ourselves almost wholly with Al( v:-.yo), minimizing this value as far as 

necessary by weighting the transmission pattern of the aperture (not the grid). 

2.5.4 Aperture Amplitude Spectrum of Circular Aperture 

The amplitude spectrum of a circular aperture of radius'a'is 

A(vjf: nal a )4(2-77 a 

(z 77 a v) (2.11) 

where J1 is the first-order Bessel function. In the NS, a = 3.56 cm and 

vg - 3.67 cm-1 . The argument of the Bessel function 217a vg = 82 when 

v is substituted for v, corresponding to the condition for D.C. .(remember 

that 

Ai(v) ^ rr aaall {au alv-vgli 

ana lv-v1 

g (2.12) 

2-18 

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because of the shift or origin to vg). 

The amplitude of the waves of J1(z) for large z is given approximately by 

Ampi. (z)..] 4 rr Z/2 

(2.13) 

When z = 82, J1(82):::0.81 in amplitude; hence 

2J1(82) . 

82 ,., 0.002 

and we conclude that the aperture "response to D.C." for an undiffused circular aperture is 0.2%. This means that if T(vr.._-; o) were 500 times as large 

as T( vt-,.fl:0, which it might well be under some circumstances, the "feedthrough" 

would equal the desired signal, leading to very serious errors in the velocity 

output. 

2.5.5 Aperture Diffusion 

To reduce the response of the aperture to the large contributions from 

o), we seek a multiplier in the spectral space (v) which is unity near 

v= v g and falls off rapidly away from v. 11 g, especially reaching small 

values as v approaches zero. The "convolution theorem" of Fourier Transforms 

tells us that multiplication by such a weighting function (attenuator) in 

v-space is equivalent to convolution of the circular aperture by a "diffusing 

function" in the physical x-space. Further, the diffusing function and the 

weighting function are Fourier Transform mates. 

For example, if we use a diffusing function 

r 2 

- 

•L (y) 7713 z 

e 2 

• -SECRETID 2-19 
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where Y is the radius and Pis the width (parametric) of the circularly symmetric diffusing function, the weighting function in v-space is 

F P = e (2.14) 

where p is the radius vector in "-space. F(P) is a function which is unity 

at the origin and falls of exponentially with P 2. 

Suppose that we wish to attenuate the "feedthrough" by a factor "d"; then 

d = e 

and (2.15) 

/3= ve 1 

If d = 1000, In d = 6.908, and P. 0.16. Since a = 3.56, /3/a = 1/22. A 

slide taken through the center of the diffusing function would drop to halfamplitude in a (radial) distance 

yl= ‘12 ln 2 

2 

=1.180 (2.16) 

After diffusion, the circular aperture transmission will fall off at the edges 

at a (midpoint) slope of slightly less than J1/217 

Note that the "attenuation" d increases very much more repaidly than /3; 

thus slightly increased diffusion provides marked decrease in the response to 

background illumination. 

2-20 

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2.6 EXTRACTING VELOCITY INFORMATION 

We have seen that the shape and make-up of the aperture largely determines the spectrum of the photosignal. Extended apertures yield narrowband photosignals at a carrier frequency dependent upon the grid wave 

frequency Vg v g. Without grid wave number v the carrier frequency g, 

goes to zero leaving a low-pass signal with bandwidth dependent on the 

aperture bandwidth and the velocity. Very small apertures yield wideband photosignals by virtue of wideband aperture spectra. These two types 

of signal, wide and narrowband, have very different noise susceptibilities, 

as illustrated in Figure 2-7 because of the received light energies (small 

for small apertures) and circuit bandwidths (large bandwidths admit proportionately large noise contributions) and different methods must be used 

to extract the desired velocity information from the photosignal. 

When the information contained in the photosignal is broadband, such 

as that produced by a flying-spot (aperture), one must resort to correlative techniques: successive cycles of the signal are compared to determine 

the shifts produced by the forward velocity Va. Since no integration (to 

speak of) occurs in generating the signal, it must be processed in the time 

domain. The aperture may be enlarged to approximate.a typical correlation 

element in the interest of higher signal without loss of information. Such 

a correlation element is the Fourier transform of the terrain power spectrum; 

to a good degree of approximation, the terrain may be considered to be made 

2-21 

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.4/04.SP 

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11.,91Mea)Z7,9412) /1-5. 7:fcceSj 

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e)/ 2E23-9/VD /1/.4 -

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Figure 2-7. Bandwidth Difference and Eoise Susceptibility of Correlative 

and Frequency Measuring Beta Systems. 

2-22 

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II 

up of small tiles, the luminance of each tile being independent of every 

other, the area of each tile being equal to the mean area covered by the 

(two-dimensional) terrain autocorrelation function. The number of such 

tiles within the field of view represents the number N of independent samples available to the correlative process of velocity estimation. 

Extended apertures, such as grids, perform an integration over the field . 

of view. When the grid is periodic, the resulting signal represents the 

harmonic component of the terrain at wave number vg, and the spectrum of 

the signal is a narrow peak at f = vg(VaVg) of width determined by the 

spectrum .of the aperture outline or envelope. Narrowband signals must be 

processed in the frequency domain. The velocity information is now contained 

in the frequency of the signal and the location of the spectral peak. 

Two ways are used to estimate the frequency of such a narrowband signal: 

counting the number of zero-crossings per unit time or correlating the 

signal with a local oscillator of known frequency (the latter being the phaselock method). A convenient hybrid of these methods correlates not the full 

wave shape, but locks the local oscillator to the square wave signal which 

results from strong limiting of the original photosignal. Such limiting is 

essential in achieving wide dynamic range. A further advantage of this hybrid 

method is that the local oscillator may be square wave also. The phase-lock 

then occurs not between the sinewaves (of slowly varying phase and amplitude) 

but between similar square waves. Simple switching and logic circuits can 

therefore be used in lieu of linear multipliers and integrators, advantages 

well known in telemetry. 

2-23 
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Although we reserve the term correlative method for image velocity 

sensing techniques which cross-correlative successive time-sequence photosignals to estimate velocity, all techniques must involve correlation in 

some form. The various techniques are generally identified by specific 

titles. However, filtering, integrating grid transmission times terrain 

luminance, matching local oscillator waveform to photosignal, etc. are all 

forms of the basic mathematical operation of correlation. 

2.7 ERROR 

2.7.1 Sources 

To devise several ways of doing the same job is often possible. This is 

evidently true in the case of image velocity systems. The differences in 

performance generally are of two types: probable error, and dynamic response. 

Usually speed of response can be traded for probable error, since both are 

factors in the basic instrumental limitation, the rate of information input 

(or the noise bandwidth of the process under observation). 

The performance of physical systems in real life is strongly conditioned 

by noise, and the sources of noise which are operative in a given situation 

determine the resultant error. 

Since terrain constitutes a random pattern in motion, it generates a noise 

which we will call pseudo velocity. Our term arose in the following way in the 

analysis of grid-aperture systems. When terrain is viewed by the photo sensor 

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through a sinusoidal grid, the photocurrent is proportional to the amplitude 

of the sinusoidal component of wave number v in the terrain within the field of 

view. If this component were coherent in successive fields of view (i.e., if_ 

the wave merely was a continuation of the original one, without change in amplitude or phase), then the photocurrent would oscillate sinusoidally at a frequency 

Va Ivg. This frequency can be interpreted as indicating a velocity Va. If the 

component is not coherent (as in a random pattern), the photocurrent wave form 

will shift in relative phase (as well as amplitude) as the forward velocity 

introduces new elements into the field of view, and the apparent frequency will 

fluctuate with these phase variations; the velocity estimate will be Vaplus the 

pseudo velocity due to the random pattern. 

The pseudo velocity is completely dependent on the aperture power spectrum. 

The purpose of the aperture (envelope) is to define the field, but its detailed 

design is motivated by the desire to minimize the pseudo velocity error. We 

have referred to this noise as terrain noise since it represents_the noise in 

the velocity signal due to the random terrain pattern. 

Terrain noise includes both amplitude and phase fluctuations of the photosignal due to terrain randomness. In strongly limited signal processing, we are' 

concerned only with phase fluctuations, or in most cases, only with zero crossings. 

However, amplitude fluctuation is important, since the question of signal drop 

out is often raised. Given an operating threshold .(Ervin) for reliable operation, 

say Emin in the zero crossing and/or phaselock modes of operation, the probability 

of dropout is given by the area of the layleich distribution 

R e- R72o? 0 (2.17) 

2-25 

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lying below R = Emin the area is 

,2 

(Prob. of _drop out) = 1-e (Emin) / "I E. 

where CY is the total photosignal power. The probability distribution for 

the duration of drop out is a more complex statistical problem involving the 

form of the signal power spectrum PE. Since Fein can be made very small in 

modern electronic circuits, a more critical consideration is probably whether 

the signal becomes lost in the noise, even though the signal still exceeds the 

threshold. 

A second source of noise is the shot effect due to discrete arrival of 

electrons and photons. For simple photosensors, such as photodiodes, the shot 

noise current is given by 

13

s 

 = 2 EBwt,o(arpperes)• (2.18) 

where E is the electronic charge, Bvjis the noise bandwidth of the signal 

channel, and to is the mean photocurrent. Current due to background illuminance plus-dark current constitutes (0 . When electron multiplication takes place, 

the shot noise is enhanced by the factor s/(s-1), where s is the dynode multiplication factor. 

Another noise contribution is due to thermal effects in the input impedance, 

other circuit components being negligible contributors because of lower impedance 

levels. The thermal or. Johnson noise is given by 

E3 = 4 KTBw (amperes) (2.19) 

-6-EeRETIO 

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where k is Boltzmann's Constant, T the absolute temperature, and R the effective 

impedance of the input circuit. Thermal noise is to be considered for semi-' 

conductor photosensors, but is not a factor in photomultipliers. 

Semiconductor noise is a contributing factor in cases involving low frequencies 

since this type of noise is inversely proportional to frequency. Carrier systems 

avoid this noise contribution by virtue of their operation at frequencies well 

above the region of semiconductor noise. Circuit noise becomes less significant 

in successive stages due to the increase in signal levels. In well designed 

circuits using modern components, it is believed that circuit noise makes a 

negligible contribution to the overall noise problem. 

The aperture may contribute signal disturbances of both random and systematic 

character. For example, random variations in grid spacing line width, or density 

will show up in the aperture power spectrum as decreased power (light loss) in 

the desired frequency component; these kinds of (draftmanship) errors do not cause 

peak spreading or distortion, and their effect is, therefore, not really significant. However, systematic errors in grid period, such as those due to grid-disc 

eccentricity, will cause frequency modulation at disc frequency. Such low-frequency 

FM can be minimized by (1) careful location of reference sensors to compensate 

for residual eccentricity, and (2) eliminating the surviving output variations 

by filtering, which, at high disc speeds, lies well above the required output 

bandwidth. 

The effects of all random noise sources can be obtained by superposing their 

corresponding power spectra: 

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1) Terrain noise has the total power and spectral distribution 

resulting from the convolution of the aperture power spectrum with the terrain 

power spectrum, as described in subsection 2.5. 

2) Shot and thermal noise contributions have the power densities 

given in equations 2.12 and 2.13 not including the Bw factor. 

The total noise power from these sources is that given in the cited equations. The spectral distribution is that of the signal channel, which is 

determined by the bandpass characteristics of the circuits employed. For example, 

in a double cascade of bandpass filters, with each simple resonant circuit of 

quality factor Q at frequency fl, the bandwidth is 

1'1 

2Q (2.20) 

and the power spectrum is of the form 

f

1 + 4Q'(f i 

(2.21) 

its 

3) Semiconductor noise and circuit noise, where significant, have 

1/f and circuit bandpass behavior, respectively. 16 general, these will notbe significant in preferred types of image velocitY sensors. 

The effects of noise are described by.different formulas for different types 

of sensors. 

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2.7.2 Correlative Systems. Such systems are unusually susceptible to noise 

because (1) signal power is quite low due to the necessary use of small apertures, 

and (2) noise bandwidth is large to accomodate the required signal information. 

Velocity error distribution for correlative systems can be assumed to be normal 

with variance 2 Crvc.; . A block diagram for a typical correlative system is 

illustrated in figure 2-8. The velocity tracking signal is derived from the 

difference of the early and late correlators. The correlation value produced 

by each of the correlators has an error associated with it which is dependent 

upon the degree of existing correlation f) and the number N of independent 

AA 

/A- qy a)Tpo7" 

Figure 2-8. Correlative System 'Block Diagram. 

2-29 

0/V4-- ci,c.z.z5 

/14". /PAX 

...4-31/ 

/A/71Z-11/17OR 2 < 

n,5-41,exkwve 
-bo 

tiles of terrain which enter into the place of continuous functions is a 

justifiable approximation which permits us to make use of tabulated values 

(from the work of F. N. Davidl and of Karl Holzinger2) of the correlation error. 

The number N is equal to the scanned arca divided by the correlation area of 

the terrain (the mean area of the Fourier transform of the terrain power 

spectrum), assuming that the correlations are performed once per scan interval 

( At ). Existing correlation f can be deduced in terms of the power 

spectra of the photosignal due to terrain (essentially the scaled terrain power 

spectrum), the power spectrum of the noise, and the early-late time shift 

± p t , a small part of. At (generally &t<1\tve(). However, in the 

noise power spectrum we must include Terrain noise due to failure to trace 

over the same tiles in consecutive scans. The basic assumption of the correlative 

method is that the signals being correlated are, hopefully, identical except 

for the displacement due to the motion in the forward direction. However, some 

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noise is introduced by the Shift of field of view, even though the scan occurs 

in a raster of lines parallel to the forward direction. If some lateral velocity 

exists, the scans will not see exactly the same sequence of tiles, and the correlation then drops off. Thus, an adequate treatment of the errors of a correlative system must take into account that the translational and vibrational effects 

on the retrace distribution. 

Let us assume that the lateral displacement of traces for the interval At 

is characterized by a distribution P6tN Then we may expect that the 

1 David, F.N., "Tables of the Ordinates -and Probability Integral of the 

Distribution of the Correlation Coefficient in Small Samples", Cambridge, England: 

The University Press, 1938. 

2Ho]zinger, K., "Tables of the Frei:able Error of the Coefficient of Correlation 

as Found by the Product Moment Method", No. XII, Cambridge University Press, 1925. 

2-30 

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average correlation of the sequence of terrain tiles will be 

PT .7-jeky)P(0,0dy. (2.22) 

where P (x,y) is the terrain correlation function. The total photosignal 

power is then 

P 2 P I Z.) ) PN. E TE IT (2.23) 

where PE 

is the power of the terrain signal and PN is the noise due to thermal 

and shot effects. Renormalizing, we obtain the signal to noise ratio 

kt Sig /noise PE 

PE( I - Pi) I -32. ' (2.24) 

which may be used to compute the resulting, reduced peak value of signal correlation (P.) . The shape of the correlation function is obtained by taking 

the Fourier transform of the combined power spectra (or the sum of the individual 

transfoims). In order to obtain the required steering signals, it is necessary to 

operate off the peak a short distance on either side, ( ZNX in displacement) 

and to take the difference of the resulting correlations. The slope of this 

difference at null is 26.4,F1/43)1Lsx. The situation is illustrated in figure 

2-9. 

• The uncertainty in the null position is times the uncertainty in 

pu, divided by the slope at null. The uncertainty in the null position 

2-31 

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.161E-VV7,-11Z-bb 

___\________ r 77 

\ / -> ../. i ..- ..," 

1 

w.E.47-4Av 

• ' 

/ 

Na Ci, 

Figure 2-9. Derivation of Steering Signals from Early-Late 

Correlations. 

(Uncertainty in null position) 

2-32 

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is also equal to. AtCiv , where CIV is the velocity error of the correlative 

type sensor. Hence, if we can find the uncertainty in WSX), we have a 

' complete method for finding the velocity error. . 

The uncertainty in evaluating a correlation coefficient is dependent upon the 

actual value of I) for the sampled population, and upon the number of independent 

samples N. The complete distribution function is given by F. N. David as 

n -2 

pz) 2 1:3.\ 2 d ...faccso 5(LE - -Y/- ) (2. 25 ) 

• -rr(n - 3)1 I ) (LICYPP 2. r-py --

where "y is the uncertain value derived by computation on the limited sample. 

Fortunately, this equation has been converted to tabular values and graphs for 

typical values, figure 2-10 shows probabilities for small N. For large N and 

small P , the uncertainty in the computed correlation is well represented by: 

p)NN (2.26) 

Knowing f)(x) from the Fourier transform of the total power spectrum scaled to 

peak value El, , we may evaluate it for , also determine the slope 

(dp/C1X) IL\ x . The velocity error is then completely computable as outlined 

above. 

The numerical values of error depend upon the particular male-up of the 

particular Beta system. 

2.7.3 Zero Crossing. Frequency measuring types of image velocity sens:Ts 

have been used for many years with only moderate success. The error evaluation 

2-33' 

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involves an examination of the errors involved in counting the zero-crossings 

of the signal waveform during a specified interval. The longer the counting 

'interval (or filter time constant), the smaller the variance of the counts 

determined. The expected number of zero-crossings is 

r 2 

f w(S)dSfw(8)d8 

• 

(2.27) 

in which Vi(8) is the power spectral density of the signal, or Ft:(8) in 

our case. The variance of No is difficult to obtain analytically. See diScussich 

by J. S. Bendat.

3 

In the absence of complete statistical results for the effect of counting time 

(or averaging time) upon the variance of the velocity, we resort to an approximation as follows. 

Consider a narrowband Gaussian noise, the photosignal resulting from a fixed 

grid. The waveform will be characterized by an autocorrelation function which 

consists of a cosine wave of frequency within an envelope which is the correlation 

function for the outline of the aperture. See figure 2-11. This correlation will 

be high (near unity) for delays close to 1-1=1/Vo_Vg slightly less at eLT1 , 

etc: If a zero-crossing (negative going, perhaps) exigtg at 1:=,() we may expect 

one near 15 ; at 7i the curve will lie 41 - R:fr Z above the zero axis, 

CY being the root mean square wave amplitude (square root power), R .the 

correlation at 7-1 , and 77 

4_ a normally distributed random variable of zero 

mean and unit variance. 

3 Bendat, J.S., "Principles and Applications of Random Noise Theory", 

'Wiley & Son; 1958. 

2-35 

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The error in zero-crossing interval is obtained by dividing the amplitude error 

'by minus the wave slope. The slope is approximately 242 1T011.i, and the 

interval becomes 

7-41 --Fr(7i) 

277NTT 

as illustrated in figure 2-12. 

Now let us determine 11 by observing the time taken for a known number 

of zero-crossings to occur in one direction. The time variance of this 

observation will be 

' 4\1:r1.71.[.-R2(T,) 

and the fractional root mean square time error is 

CT x A-WRV!) 

T _SAW 11 NT 21/2 n2 

7r 

--R1(7)) 

2NT (2.28) 

The fractional velocity error is the same quantity. 

In carrying through this approximation for velocity error due to zero counting 

we have not counted zeros per given time, but time per given zeros. The troubles 

which arise in statistical theory due to the discreteness of zero-crossings do not 

occur in physical systems used in the Beta system. This is because such systems 

operate, generally, in terms of time rather than count, and our treatment' is consistent with this distinction. 

The results expressed in (2.28) are expected to be low by the factor 174. 

.................. 

2-36 _smftErter 

1 
2-37 

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/ 

/ 

/ 

N 

\ 

\ 

\ 

Viol 

I 

/ 

/ 

7 

2.,-11 

\ SLOPE \ 

2.7r0' ' "I.. \ 

f I 

\ 

/ 

/ 

Cij ri -- R 2(ri ) 

. 

\ 

/ 

\ / 711 1/I. R2(4 

\ / 4 17Pri- 

%.... ...... 

I I 

0 

P057 

Random-Time Error Derivation of. Zero Crossings. ('or bandwidth limited 

noise in terms of normalized autocorrelation function of photosensor 

• 

signal. Zv is a normally distributed noise with zero mean and unit 

I, I 

variance, having uniform power spectral dibtribution for-2t <T<+ 2tI) I— — 

otherwise zero.) 

Figure 2-12. 

I 

ENVELOPE EQUALS 

AUT000RRE LATI 0 N 

/

FUNCTION OF APERTURE 

ScALto BY Ittra_ 

RE ('r) 

0 

• 

NORMALIZED AUT0CoRRELATION FUNCTION 

P056 

. COS 2173

/1 TIMES ENVELOPE 

Figure 2-11; ,Autocorrelation Function of Photosensor Output Signal E(t) for Rec-. 

tangular Aperture width V/ and velocity Va .11=1/{vg(vcivg)].) 
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The reason is that we have ignored the fluctuation of the amplitude of the 

photosignal wave, setting it equal to a.Cr . A more rigorous treatment, 

which includes the amplitude, has yielded the larger result. 

The error in the frequcneing measuring case is thus shown to be dependent upon 

R(T) and NT T. R itself is the autocorrelation function relating to 

the aperture envelope. The term 7-1 1=j/(/(11, may be quite large when \/c6 is 

small. In addition, for a given interval of averaging, N Twill decrease as 

VCU does. The result, which depends in detail upon the shape of R(T) is 

essentially that the fractional velocity error is inversely proportional to Va, 

Such a reasonable outcome conceals the fact that the timing process utilizes 

multivibrator waveforms, in conventional devices, which require long time constant 

filtering to avoid noisy velocity outputs. 

2.7.4 Phase Lock. A phase-lock loop makes use of a null-correlator or phase . 

error detector and a local oscillator which can be phase modulated in proportion 

to the phase error derived from the detector. Many ways of implementing the - 

phase-lock principle exist, but most telemetry and instrumentation applications of 

phase-lock use strongly limited waveforms which look like square waves of nonuniform period. The local oscillator then takes the form of a voltage-variable 

astable multivibrator which generates a square wave of frequency proportional to. 

the. phase error'voltage. This is the usual FM phase-lock method. 

A second method uses the same type of phase comparison, but uses a phaseshifted, fixed-frequency, reference oscillator in lieu of the variable frequency 

oscillator. The fixed frequency wave is then shifted forward or back in *phase at 

a rate sufficient to keep step with the waVe being tracked. The voltage applied 

to the phase shifter is the integral of the phase error voltage, which remains 

-Rr7eRET113- 

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proportional to phase rate and, therefore, to velocity. The advantage of the 

phase-shifting technique is that relative velocities can be determined;' that is, 

the reference frequency, which is phase shifted,- need not be constant. This 

allows compensation for unwanted signal variations, such as the elimination of 

tape wow in the magnetic tape recording of telemetry records. 

The error in a phase-lock loop depends upon the averaging time employed in 

the phase comparator and upon the significance of various spectral components of 

the signal. Our interest will be primarily in a phase-lock method in which the 

signal bandwidth is very much less than the bandwidth of the phase-lock loop, so 

the filtering is explicitly associated with the output filters and not with the 

loop. .We are then concerned only with the fluctuations which survive the output 

filter. 

The output of the phase tracking system is the rate of shift necessary to 

keep up with the input waveform. The variance in this phase rate may be computed 

from the zero-crossing interval 71 and the probability for phase fluctuations, 

46 4) , given by- 

_ 1)2(T) z-3/2 

Mocb) 27r ( I -le) -7215:-"I (2.29) 

where [3 p(T)cos A and f)(11) is the signal correlation function. 

On the assumption of Gaussian form for NNE)) where 1)(7 ) is nearly unity, 

we obtain a phase variance 

2[1 

71-137 (7-1) --- 

(2.30) 

2-39 

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and a consequent velocity variance 

cr". 2-rr vg-ri ) 2[1—p(-;)] (2.31) 

This variance represents the total noise power in the output prior to filtering. 

The noise is uniformly distributed over a total bandwidth of f1 = (Va+ Vg) vg. 

An output filter of low-pass bandwidth Bwo will then show an output velocity 

variance Bwo/f, times that of equation (2.31). 

2.7.5 Analysis 

Figure 2-13 illustrates the factors and relationships involved in transforming the terrain brightness B into the photocurrent I. The terrain brightness B (x,y) when imaged through a lens of f-number N, optical efficiency E:, 

and collective transmission factors V, produces an image illumination 

EVB(x,y)/4N2 foot candles. If the effective aperture, including the spatial 

effects of grid transmission, is A(x,y), the integrated photocurrent is: 

EVS1 

= 4N2 f f (x,y) A (x,y) dxdy amperes (2.32) 

aperture 

where we have used the same coordinates for the terrain and its image, 

mapping one upon the other using an unspecified magnification. The magnification is significant only in relating the grid period to a corresponding 

period in the terrain pattern. 

As the optical system of Figure 2-13 is translated with respect to the 

terrain, the photocurrent fluctuates about its mean, io. For a 50% opaque 

grid, the D.C. photocurrent is 

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777W )//r 2W/C117- 

E46 ,w or .6 NO ER43 

• 

TErw,57,7ili 

/rOD r 

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2-IN 

• . 

04= piieriG•f•/}/1/91w7 --aMt.S.6,27 7 O./V 

4AfP/R1_55` 

Figure 2-13.Illustrating the dependence of photocurrent I 

upon terrain brightness B and optical parameters. 

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C o E Vg/Ag 

8 N 2 

amperes (2.33) 

where A is the entire area of the aperture. Note that the average image 

illumination does not involve magnification. 

The fluctuation of the photocurrent will have a mean square value PT. 

F VSIf `:"" 

11-N 2 J PA61, P)P13(val,)civclu (2.34) 

where PA( v, IL) is the power spectrum of the aperture and P13( v, A ) is the 

power spectrum of the terrain luminance. 

This simple relation between PT, PA, and Ph holds even for the moving 

grid-moving aperture situation provided PA is the power spectrum of the 

ni t t.

dynamic aperture, itt).r , projected on the ( v , u ) plane. 

2.7.6 Velocity Error 

The effect of light level enters through the noise power, pN, which is the 

sum of the photocurrent mean square fluctuation due to thermal and shot effects:_ 

14-R1-} 

R. OW (2.35) 

where Bi4 is the noise bandwidth of the signal channel. 

The variance (mean square error) of the velocity determination by the Beta 

System is 

(2.36) 

2-42 
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where vg is the grid wave number, 1-1 is the period of the carrier frequency 

and P is the photocurrent normalized autocorrelation for lag 11. 

The autocorrelation P(r1) is easily shown to be the weighted average of the 

autocorrelations for the terrain signal and the noise 

• p PT PTI fr ) Ph PI4(ri) 

Pr + PN (2.37) 

The noise autocorrelation is simply that associated with the filters which 

limit the signal channel. The terrain correlation is obtained as the convolution of the correlation functions of the aperture and the terrain. 

Because r1 is the period' of the carrier signal produced by the grid motion, 

there is little evidence of grid motion in the formulae, since only the envelope 

of the correlation function of aperture is used. 

2.7.7 Correlation Functions 

The signal channel of the Beta System is limited in frequency by a cascade 

of two single-resonant filters, each of the form, relative to fl, the carrier 

frequency 

PF (c) --- 

single filter spectra density, 

1+ 

whence Rp 7% e--2-rrfoir),_ single filter autocorrelation 

2 

The correlation for the dual. (cascaded) filter is obtained by convolving 

two single-filter correlations: 

—artcc, ri 

R FFt(r) 2 { 4-2_77 -F,,r} 

SECRCIP 

2-113 

(2.38) 
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from which the normalized correlation is found to be, for small T , 

approximately 

efo2 r 2 ) • (2.39) 

The autocorrelation for the aperture will depend, in envelope, only upon 

the shape and transmission weighting of the aperture without the grid. A 

circular aperture of radius R and uniform transmission has a correlation 

C ( X ) 

C (A) = R2 (2 q' - sin 2 IF ) (2.40) 

where A= 2 Rcos 111.. 

The variable A is the displacement variable; it is related to r by the 

aperture velocity Va; i.e. X= Va T. For small values of ri, with which we 

are concerned, 

C (A > 0) () a A 

rr r\ (2.41) 

The desired signal correlation would then be as follows (assuming that the 

terrain spectrum is much broader than the aperture spectrum): 

gyo,r\ (2.42) 

/ 

A better Beta System is obtained if the edges of the aperture are shaded or 

diffused. We may consider that the diffused aperture AD is formed by convolving 

the original sharp edged aperture A by a diffusing function in the form of a 

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two-dimensional Gaussian probability function D (r). 

ra 

e apa • 

a. n 13 2- 

(2.43) 

Here 0 characterizes the width of the diffuser. An approximate result, 

good where A is small relative to 0, is 

Aa 

cp (A) ITR2- ITTIBF 

from which one obtains 

Va,?' ra 

pf (7.) :I { I Li- R 

(2.44) 

(2.45) 

The advantage of the diffusion lies in the fact that p(r) for the diffused 

aperture falls off quadratically while the undiffused correlation drops linearly 

with r. 

Whenever 

„e

r1Z vct, or 7;-R R— p R loR TIT (2.46) 

The diffusion is advantageous when 80>Y2r,57, within the range of validity 

of the approximation CD. 

2.7.8 General Error Formula 

Having all the parts derived in previous sections, it is, a simple matter 

to put these parts together into one complete formula for the velocity variance 

n- 2 -viz. ., 2 RM 

) 

V D; ri...5:r , 2 r, a t 

a t ..:: (712:, ( 1 )(1-1' ) IT13 i\% - 1 77 1" 4-77 -to riPI-1 (2.4.7) 

vel, • {:i ?,7 7 ' 1.0 r, n PT 1- Fri 

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vc,2- Pr 

= RFT 477 fc) 

+1 a n PT PN 

BIF -005 -112 -68 

2.48) 

When Va 

o, the limiting velocity variance is 

Va, o 4-fy, fat

fl 1,92- CP-v-1- (2.49) 

In these formulae, fxis the corner frequency of a single-pole output 

filter. The dimensions of vg are consistent with those desired for Va. 

It is significant that the variance contains no contribution from PIT 

when Va-> o, but a large PT(as a result of good luminance levels) decreases 

the variance due to Pnby the factor (PT+ PN)/Bu. 

2.8 COMPARISON OF METHODS 

We come now to a comparison of three types of image velocity sensing 

systems: frequency measuring, delay correlative, and phase-rate measuring. 

The correlative method has small aperture and wide bandwidth. The others are 

large aperture narrowband systems. 

While the wideband correlative method appears to make the most inclusive 

use of terrain power, it does so only at the expense of large noise acceptance. 

Were the signal power quite large, the noise penalty might be acceptable. 

But, in low-light situations the noise can easily swamp the needed signal 

and make velocity errors intolerably large. An aperture dimension will exist 

for which the improvement in light level just offsets the reduction in number 

of samples. This represents the minimum error achievable by the phase-rate 

system. 

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Such television-type tubes as the vidicon are capable of accumulating charges 

during exposure to light and releasing the charges when a scanning beam falls 

upon the individual areas. By virtue of this capability for integration, it 

would seem possible to achieve signal levels associated with maximum area 

apertures while using the full bandwidth of the terrain. But the.noise bandwidth is still very much larger than that of possible narrowband systems by, 

at least a factor of 1000. 

The frequency measuring method employs a full yield aperture, thus 

acquiring almost the maximum signal levels achievable. However, since the 

frequency of the photosignal is f = lityg, the frequency is too low to afford 

the necessary accuracy together with adequate dynamic response. And, as we 

have noted earlier, the fixed-grid ou6put fails entirely at zero velocity and 

below. 

The phase-rate sensor retains the advantages of the frequency method: 

a) high signal level 

b) low noise bandwidth 

but removes the problems of velocity ambiguity by usual carrier operation, 

which provides: 

c) essentially unlimited range of velocity 

d) high response speed. 

These advantages are obtained without penalty in cost, complexity, size 

or weight. 

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Section 3 

DESIGN APPROACH 

3.1 GENERAL DESCRIPTION 

The Hycon Beta System, Model HG-469B, is a continuous analog device, 

capable of tracking random image motion having frequency components 

up to several hundred cycles per second with peak velocities of several 

inches per second. Because of the device's unique acquisition capability, 

no external input signals indicating initial motion direction are 

necessary, and there is no possibility of ambiguous polarity of output 

voltage versus input image motion direction. 

The HG-469B is designed to sense and measure image velocity at the 

center of the format of the image provided and resolve it into two vector 

components along perpendicular axes. The HG-469B operates on the 

heterodyne principle and utilizes phase-rate measurement to determine 

image velocity. Its output is in the form of two unipolar analog voltages, 

one representing x and the other y. The magnitude of each voltage is 

proportional to its respective image velocity component, and the 

polarity of each voltage (separately supplied) is indicative of the 

direction of its corresponding image component. The full scale design range 

is ±0.3 inch/sec (referred to the input image plane) for each orthogonal 

component, and the corresponding full scale output voltages are +5.0 volts 

for each component. Frequency response is that of a single lag located at 

1 Hz plus a notch filter at 30 Hz. 

The.system consists of two major components, a detector assembly and an 

electronics package assembly. 

I 

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3.2 DETECTOR 

The detector, shown schematically in figure 3-1 contains optical components 

for re-imaging a 2.8 inch diameter input format to 0.28 inch diameter. It also 

may contain- a lamp array to provide a self-test capability. 

3.2.1 Optical Subassembly. The 2.8 inch field will be formed by an f/7.0 

system. The relayed images will lie in a plane parallel to the input object 

plane at a distance of 8.o inches from it. 

The two optical channels will be formed by an achromatic beam splitting 

surface. 

The rays from the lens are divided by the first beam splitter into a vertical 

and a horizontal beam. The vertical beam is reflected by a series of 3 mirrors 

into a relay lens. The f/0.636 relay lens will cover a 102° field of view. The 

lens will be designed as a low resolution/high contrast lens. The horizontal 

beam from the first beam splitter is reflected at 90° down to two penta-reflectors 

and out at right angles into a second relay lens. Both paths have exactly the 

same total length to a common focal plane. The lower image is inverted but not 

reverted and the upper image is rotated exactly 90° with respect to the lower 

image. 

3.2.2 Chopper Disc. This disc is a 4 inch diameter optical flat upon which 

has been deposited a precise pattern of alternate transparent and opaque sectors 

having equal angular dimensions. For the image scale condition existing after a 

reduction of 10:1 at the prime location, a disc having 1024 spatial cycles is 

optimum. Theory indicates that the grid clear dimension should correspond to the 

terrain correlation distance. Stated in other words, the statistical terrain 

particle diameter should be equal to the width of the grid clear zone. The mean 

grid clear zone width is approximately 0.005 inches, which when referenced to the 

earth at operational ranges represents terrain distances of 40 to 140 feet. The 

grid wave number, %, is 92 cycles/inch. 

3-2 tfels.it—Q—ef 
4693-2 1:3IF-°°5-112-68 

Volunic_, II 

X PROTOMULT I PLIER 

Y RELAY LENS 

CC 

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Lai C.7 

u- co") 

C:= O 

2 

ocC 

g= 

LsJ 

O. LIPage G 7 

rt 

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3.2.3 Aperture Shading. The physical dimensions of the image sensing 

aperture are established by an aperture mask located either between the lens 

and the front surface of the chopper disc or in front of the field lens. When 

located in either area the mask is deliberately spaced so that the mask shadow 

has unsharp edges. This edge gradient diffusion is referred to as "aperture 

shading". 

In an image stabilizing system such as the HG-469B, there is usually a 

transverse image velocity component existing in each aperture in addition to 

the primary image velocity component. The optimum aperture shape is circular, 

since an aperture with circular symmetry will show no change in correlation 

with transverse velocity. Aperture "shading" or edge gradient diffusion 

minimizes the effects of finite aperture diameter. Shading provides a reduced 

weighting for the power contributions of image elements which are in the process 

of just entering or leaving the image aperture thereby minimizing perturbations 

due to spectral reflections and isolated "hot spots". 

3.2.4 Detector Electronics. Output light from each of the light 

collecting systems is supplied to the x and y photomultipliers. The output 

signal from each of the photomultipliers is fed to its respective preamplifier. 

The output signal of these image preamplifiers is a carrier frequency of 

30,720 Hz which is phase modulated by the ensemble of image elements. Detection 

of the rate of change of image carrier phase relative to a reference phase 

constitutes the principle of image velocity determination. 

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3-4 

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Means for generating separate x and y reference signals is provided by a 

pair of stationary reference light sources located on each side of the x and 

y image apertures. The light from these four reference sources is chopped by 

the rotating disc and the resulting A.C. light signal from each pair is received 

by its respective photodetector. The electrical output signals from each pair 

of detectors are summed and fed to their respective reference preamplifiers. 

Automatic gain control is provided to the photomultipliers by varying the 

high voltage. The signal sensed for this purpose is the D.C. current of the 

"x" photomultiplier. 

In summary, the detector portion contains the optical, mechanical and electrical 

components for generating and preamplifying four basic signals, x image carrier, 

x reference carrier, y image carrier and y reference carrier. These signals are 

v,,,plied to the electronics package through a single cable having a length of 

10 feet. 

3.3 ELECTRONICS PACKAGE 

The electronics package contains circuitry for generating all of the required 

outputs in response to the four basic detector signals. The functional split 

between the detector and the electronics package inherently results in most of the 

thermal dissipation occurring in the electronics package. As shown in figure 3-2, 

the following functional circuits are contained in the electronics package: 

a. Four limiters, one for each of the basic detector output signals. 

b. Four filters. 

c. Digital tachometer and disc servo amplifier. 

d. X and Y phase comparators. 

e. X and Y phase modulators with associated integrators, gate generators, 

and output filters. 

f. X and Y signal correlators. 

g. Low voltage power supply. 

h. CFE A/D converter. 

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3-5 
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3.4 DESIGN DATA 

This covers the broad design concept and detailed circuit design chosen to 

insure compliance with all aspects of the design specifications. Since the 

proposed Beta System is so similar to Hycon's HG-453 production sensor, frequent 

reference is made to experimental data obtained during the course of development 

and testing of the HG-453 device. 

1 

The point of departure is the generalized formulation expressing proportional 

velocity error in terms of the basic system parameters. 

This formula is: 

fir A 0 + 41r a-F0 go* 

28(8D-gg) 

AoM 2-1- 130 

(3.1) 

For the Beta System, the various symbols are defined as follows: 

Va= the input image velocity in inches per second 

crVa= the r.m.s. error value of indicated velocity in inches per second 

vg= the number of complete black and white grid cycles per inch which is 100 

fx= the lag "corner frequency" which is 1.0 Hz 

fl = the carrier frequency which is 30,720 Hz, this is the product of the total 

number of black and white grid cycles (1024) and the rotational speed 

(30 rotations per second) 

fo = the half bandwidth of the carrier bandpass filter and is equal to fl 

2Q 

For this system fl = 30,720 and Q - 1000, therefore fo = 15.36 Hz 

Bo= the mean square noise current due to shot and thermal effects 

1 This formulation is developed in a

• 

 special report by 

Dr. Philip A. Shaffer Jr. (5 January 1967). 

3-7 

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Ao = the mean square signal current at the carrier frequency 

M = the modulation factor 

D = the diameter of the circular image aperture, which is 0.28 inches 

(3 = the dimension of edge diffusion of aperture, which is 0.014 inches 

We rewrite the above formula for the case where Va = 0, yielding the 

following simplified result which shows the r.m.s. null uncertainty: 

Va (Va.= 0

)=Ifl fx/Bo 

Q1)9 A0M2+Bo 

(3.2) 

From inspection of the above two formulas, it is apparent (and fortunate) 

• 

that the optimum parameters in formula (3.1) resulting in the minimum value 

also results in the minimum value of r.m.s. null uncertainty 

Va 

in formula (3.2). 

With these fundamental relationships- defined, the following paragraphs 

reasoning used in establishing the key design parameters. 

3.4.1 Optical Scale, Disc Geometry, and Rotational Velocity. On the 

basis of experiments conducted by Hycon in the course of developing the HG-453 

V/H sensor, it was found that spatial filters used in narrow band devices should 

be designed to look at terrain "cells" of 30 to 200 feet in diameter for terrain 

in general. This is in general agreement with the analysis of a large sample of 

terrain imagery. 

When the image at the focal plane of the prime system is optically reduced by 

ten diameters in the process of being relayed to the Beta Sensor disc, the resulting 

image scale is then fully compatible with the 4-inch diameter 1024-cycle disc. 

The nominal grid clear sector width in the sensing aperture is approximately 

.005 inches, which at a minimum operational slant range covers 40 feet of terrain 

and at a maximum operational slant range covers 140 feet of terrain. 

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The choice of disc r.p.m. involves both the fxand fl terms in equations 

(3.1) and (3.2). Due to the effects of minor eccentricity in the mounting 

of the disc relative to the motor shaft, a small but measurable sinusoidal 

noise signal is generated having a frequency: 

fete. 

RPM (3.3) 

The disc should turn at a speed which will cause the disc eccentricity 

signal to fall at a frequency which is appreciably higher than the "corner 

frequency" of the single lag. Since in this system the single lag is placed 

at 6.3 radians/second or 1 Hz, a choice of 30 Hz/second is suitable. This is 

obtained by turning the disc at 1800 r.p.m. The resulting carrier frequency, 

I'D is 30,720 Hz which is a very practical operating frequency for signal 

processing structures such as the filters and phase modulator multivibrators. Equation (3.1) shows that a small ratio of fx to f1 is desirable 

to minimize indicated velocity error, and in this case: 

fx/fi = 1 = 0.0000326 

30,720 

3.4.2 Disc Mounting Eccentricity Analysis. Figure 3-3 shows how the formula 

expressing L\fc, the peak carrier frequency deviation, was derived. This formula 

is: 

Afc= e/r fc 

where e is the eccentricity in inches, r is the disc radius, and fc is the carrier 

frequency. 

Since the method of signal processing employed responds only to the difference 

frequency between image and reference carriers, no eccentricity signal results 

when the image and reference apertures are at coincident locations relative to 

the disc. Thus the reference and image apertures are placed as close together 

SLLOVZi.:171D 3_9 

SECRETED Volume II 
AXIS OF ROTATION AT 

ANGULAR VELOCITY W 

MAX

r 

CONDITION 

PHOTO 

DETECTOR 

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SYMBOLOGY 

1.31x -vv.) -i1C....UV 

. 1 . g - Angle subtended by one disc cycle (radians). 

-.Angular velocity of disc shaft (radians/second). 

r -Disc radius (inches). 

>v-Arc wavelength observed by point receiver, Rx(inches). 

.3 -Disc velocity (inches/second) observed by point receiver. 

T -Temporal period (seconds) observed by point receiver. 

= T - -6 1 

S = (r-e)(.0. 

Xvia-x---(r-ve )0 and 

X min 

(r-e)W - 

X max 

(r-e)W = 

Tmin = 

T max. 

X rnin=(r-e )0 

(r-e)(9 . 

Tr-al ) IT1 a3C 0 

(r+e) lJ (r-e)(4) 

(r-e) (rie)0 (position shown in sketch 

The peak-to-peak carrier frequency deviation is 

. = 

- (r-e)W 2 A f0f f - c%.,12 

max min r 

f

c 0 (r-te)e 

or n f e , the peak, carrier deviation = f 

r e• 

Fl.iwre 3-3. Disc Eccentricity Analysis 

r" 3-10 p/ 

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as possible, at identical disc radii, to minimize differential frequency deviation. The peak differential frequency deviation produced by image and reference 

apertures separated by disc angle 0 is: 

fdiff = e/r fc (1-cos 0) 

An order of magnitude reduction in this peak differential frequency deviation 

is obtained by situating two reference generators symmetrically about the image 

aperture and summing their outputs. 

Computing the magnitude of the 30 Hz eccentricity noise for the proposed system 

having the following properties: 

e = eccentricity = 1 x 10-4 inches 

r = radius = 1.75 inches 

fc = carrier frequency = 30,720 Hz 

95 = angular separation of image and reference apertures centers = 8° 

fdiff . 0.1 e/r fc (1-cos 0) 

= 0.1 10-4 (3.07 x 10) (.01) (3.6) 

1.75 

= 1.75 x 10-3 Hz 

The resulting root mean square velocity error due to eccentricity, Vae, referred 

to the disc image scale is expressed by 

O Vae= 0.707 A fdiff 1/1,2 

. 1 (fn) 

( 3 . 7) 

;7 

Where fnis the eccentricity frequency (30 Hz) and fxis the corner frequency 

of the single lag filter (1 Hz). 

Substituting, we have 

a Vae- = 0.707 x 1.75 x 10-3 x 1 x 10-2 x 1 

1 +j30 

= 4.1 x 10-7 inches/second 

3-11 Pays--- 
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Referred to the primary image plane, aVae is ten times as large, or 

4.1 x 10-6 inches/second which is negligible compared to the total allowable 

null uncertainty of 0.01 inches/second. 

3.4.3 Choice of Operating Bandwidth. It is evident from equation (3.2) 

that the null error of the HO-469B is highly dependent of the Q 'factor of the 

carrier bandpass filters. In equation (3.2), Q appears in the denominator, 

Do is inversely proportional to Q, and the quantity A0M2 is large compared to 

Bo. 

This results in aVabeing approximately proportional to Q-3/2. The maximum 

value of Q, consistent with the requirements of signal bandwidth and carrier 

frequency stability, should therefore be employed. 

The required signal bandwidth for a frequency modulated wave (see footnote 1) 

depends on both the frequency deviation, Afar , and the sinusoidal modulating 

frequency fm. The ratio of Afl to fmis defined as the modulation index, 

The value of 0 determines the number of significant sideband pairs and this 

number can be calculated by the use of Bessel functions. Or more simply, a useful 

rule is that a frequency-modulated wave contains sideband components of importance 

on either side of the carrier wave over a frequency interval approximating the 

sum of the frequency deviation and the modulating frequency. 

When the instantaneous frequency of a frequency-modulated wave is varied in 

a more complex manner than that corresponding to sinusoidal modulation, the 

frequency spectrum becomes very complicated. The sideband frequencies present 

include not only those that are obtained with each modulation frequency acting 

separately, but also various combination frequencies. However, although complex 

modulation greatly increases the number of frequency components present, it does 

not widen the frequency band occupied by the energy of the wave. To a first 

3-12 

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approximation, this band is still approximately twice the sum of the maximum 

frequency deviation at the peak of the modulation cycle plus the highest modulating frequency involved. 

To determine the maximum carrier frequency deviation for the HG-469B, the 

following formula is used: 

• 

fi max - 0.1 Si max . Vg ( 3 .8) 

where the factor 0.1 accounts for the 10:1 optical reduction between the primary 

image plane and the disc plane, Si max is the maximum image velocity in inches 

per second, and vg is the grid wave number in cycles per inch. 

Substituting numerical values, we get: 

Afi max = 0.1 (0.5) (100) 

= 5 Hz 

The required signal bandwidth is: 

BW0 = 2(5 1) = 12 Hz 

This bandwidth figure is very conservative because it assumes that the entire 

peak carrier deviation is due to a modulating frequency of 1 Hz. 

An additional consideration involved in the choice of operating bandwidth 

is carrier frequency stability. In the HG-469B, carrier frequency is equal 

to the product of the total number of disc spatial cycles and the number of 

disc rotations per second. This causes the carrier frequency to vary directly 

as a function of disc speed, hence a description of the method used to servo 

control the disc speed and an error analysis will be given. 

3.4.4 Disc Servo. Figure 3-4 is a functional block diagram of the disc 

servo. As shown in the upper left hand corner of figure 3-4, stationary 

reference illumination is provided on each side of the image aperture by a 

pair of gallium arsenide electroluminiscent diodes. Each diode radiates 

approximately twenty-five microwatts at a wavelength of 0.9 microns. The 

3-13 

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Volume II 

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t.1 

Faye ..... 
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chopped reference light emanating from the rear surface of the disc is collected 

by a pair of reference photodetectors. The resulting electrical outputs of 

these photodetectors are summed into the input of the x reference preamp. The 

output signal of the preamp has a frequency of 30,720 Hz when the disc shaft 

is turning at 30 rotations per second, the design center value. If the shaft 

speed is more or less than 1800 RPM, the carrier frequency is correspondingly 

higher or lower than 30,720 Hz. 

In order to achieve a carrier frequency stability of 100 parts per million, 

or .01%, a coarse-fine type of error sensing is employed. A "coarse" digital 

tachometer (frequency-to-D.C. Converter) having a long-term stability of approximately 0.5% generates a D.C. voltage which is proportional to input frequency. 

This voltage is compared to a stable D.C. reference voltage corresponding to 

the desired reference frequency of 30,720 Hz. The difference between the tachometer output voltage and the D.C. reference voltage constitutes the coarse servo 

error signal which is fed into the servo compensation network. The output of 

the compensation network is chopped, amplified, and fed to the control phase 

of the disc servo motor which results in the reference frequency and servo shaft 

speed "pulling-in" to within 1% of design center value. The resulting frequency 

is then 30,720 Hz 11% which is fed into the mixer. A precise 30 Hz frequency 

11.001%) generated by a crystal oscillator is supplied as a second input, fo, 

to the mixer. As a result of the mixing or modulation process, frequencies 

corresponding to the sum and difference between frand foare generated. The 

sum frequency of approximately 60 KHz is discarded by a low pass filter connected 

in series with the mixer output leaving the difference frequency (fr - fo) of 

720 Hz 1.300 Hz. This frequency is supplied as input to the fine digital tachometer. The fine digital tachometer produces an output D.C. voltage which is 

proportional to input frequency, and supplies this voltage as input to the 

fi 

L. 3-15 Page 
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coarse-fine selector structure where it is compared to a precise D.C. reference 

voltage corresponding to 720 Hz. The difference between the fine tachometer 

voltage and the D.C. reference voltage constitutes the fine error signal. A 

logic circuit in the coarse-fine selector transfers the loop from the "coarse" 

to the "fine"mode at the instant the coarse error signal drops below 1%. In 

"fine" mode the disc speed and carrier frequency are locked to within .01% of 

design center value. This results in a reference carrier frequency of 30,720 

13 Hz. 

3.4.5 Bandpass Filters. The image channel signal to noise ratio is 

dependent upon the noise bandwidth. Accuracy is enhanced, especially at low 

light levels, by restricting channel bandwidth as fax as possible. The choice 

of channel bandwidth involves consideration of both signal bandwidth and carrier 

frequency stability. 

The total bandwidth is equal to twice the sum of the signal bandwidth and 

the carrier deviation due to disc servo stability or: 

BW0 = 2(5 + 1 + 3) = 18 Hz 

The required bandpass filter Q in terms of operating bandwidth and carrier 

frequency is: 

fc Q = B140 (3.9) 

Substituting numerical values we get: 

' 

30,720 

18 = 1770 

It is important that identical phase responses be maintained in reference and 

image bandpass channels since any differential phase deviation rate between 

these signals is interpreted by the processing circuitry as a velocity. This 

consideration dictates the use of matched filters in both the image and reference 

channels. 

3-16 

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3.4.6 Limiters. The use of hard limiting in the image and reference 

carrier channels (following the bandpass filters) enables the sensor to 

operate effectively over an extremely large range of terrain luminance. The 

limiters, or zero crossing detectors, are of the type which utilizes diode 

feedback around a high gain operational amplifier. This infinite clipping of 

the signal virtually eliminates phase distortion caused by amplitude fluctuations 

and noise. Other types of limiters such as the Schmitt trigger'and multistage 

limiter clipping circuits can exhibit large errors in zero-crossing information. 

The Schmitt trigger introduces distortion due to its large on-off hysteresis 

characteristic. The circuit does not change state when the input signal passes 

through zero but at different levels for positive-going and negative-going 

signals. 

Distortion in multistage limiters is caused by a shift in the average 

reference level after each successive stage of limiting. The shift in average 

level arises because the signal cannot be fully clipped in one operation. Large 

peaks (which may be of one polarity only) are clipped by the first stages of 

the limiter, thus altering the waveform to produce a different average level. 

The following limiter stage then clips the resultant signal about the new 

average level, which can completely change the character of the original signal. 

The large voltage gain and dynamic range of the diode feedback limiters used 

in the HO-469B permit full clipping of reference and image carrier signals by a 

single stage of processing. 

This results in precise determination of zero crossing information and maximum 

accuracy in subsequent processing circuitry. 

3.4.7 Phase Comparator. In the HG-469B,sensor, image angular velocity is 

obtained in terms of the rate of shift of phase of the reference wave needed 

3-17 

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to keep in step with the terrain photosignal. Both signals are amplified, 

limited, and converted to sequences of pulses which are used to trigger the 

phase comparator and the phase modulator. The phase comparator is, in reality, 

a flip-flop which is set for each terrain signal pulse and is toggled for each 

reference pulse. The average output voltage of the flip-flop yields a phase 

error signal whose magnitude is proportional to the phase difference between 

reference and terrain zero crossings, and whose polarity is indicative of 

relative lead or lag. 

This type of comparator is known as a sawtooth phase comparator2. The 

use of the sawtooth phase comparator as the error sensing element in the phaselock loop of the HG-469B results in superior acquisition and tracking performance 

because its output is linear for phase differences of plus and minus 180 degrees. 

In the phase-lock loop, loop gain is the product of the gains of the phase comparator, integrator and phase modulator. The dependence of overall loop performance 

on loop gain is of prime importance. For a specified noise bandwidth, a phaselock loop having a damping factor of 0.5 yields optimum acquisition performance. 

Acquisition is defined as locking the output phase of the reference phase modulator onto the phase of the terrain signal that is perturbed by noise. The 

range of damping factors from 0.5 to 0.85 yields near optimum performance while 

decreasing the damping factor below 0.5 causes rapidly deteriorating performance. 

The effects of a drop in loop gain are significant in two respects. First, a 

lower gain results in decreased noise bandwidth. Secondly, the loop damping 

factor and natural frequency are adversely affected by a low loop gain. The 

overall effect of a nonlinear phase comparator characteristic would be a decreased 

2

Bell System Technical Journal, March 1962 

3-18 

Page...... 

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ability to acquire and track terrain signals over a wide range of light levels. 

3.4.8 Phase Modulator. The output signal of the sawtooth phase comparator 

is fed into a loop compensation amplifier having proportional plus integral 

transfer characteristics. The output of the loop compensation amplifier supplies 

input to both the phase-lock loop integrator and the output filter amplifier. 

This input signal to the integrator constitutes the wide band analog of the rate 

of change in the phase difference between reference and image carriers. Output 

voltage from the integrator is used to control the phase modulator. 

The phase modulator consists of a string of seven voltage-variable delay 

multivibrators. Phase is retarded (with respect to the reference) by increasing 

the delay interval of each member of the string. Ultimately, the string will have 

been extended or collapsed to its fullest extent. When one of these conditions 

is sensed, the whole string is reset a multiple number of waves to its midrange 

delay, ready to continue with the process. The voltage which drives the delay 

string is not the comparator output but the integral of it. Any phase error 

will cause the phase shift to grow until the phase error is nulled. The speed 

of the phase-lock loop is quite high, as allowed by the high operating frequency, 

hence, the loop is capable of following velocity fluctuation to the order of 

1000 Hz, well beyond the required limits. 

3.4.9 Signal Correlator. In any type of instrument designed for measuring 

image velocity, a built-in provision for monitoring terrain signal quality and 

tracking loop performance is highly desirable. A monitor signal can be 

effectively used in the sensor to maximize the accuracy of the sensor 

output voltage. 

The HG-1469B sensor employs a cross-correlator to monitor the degree of 

coherence between the image carrier signal and the phase modulated reference 

3-19 

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signal existing at the output of the phase modulator chain. The output of 

this cross-correlator, or signal correlator, is at a predetermined maximum 

value when the following two conditions exist: 

1. The received image carrier signal displays a signal-to-noisef ratio 

equal to or greater than approximately 20 db. 

2. The phase-tracking loop is tightly locked on with a phase error of 

less than 5 degrees. 

In the event that either the image carrier signal-to-noise ratio decreases, 

or the tracking performance of the phase-lock loop drops, the resultant output 

voltage of the signal correlator is reduced. The signal correlator output 

voltage is supplied to an amplitude comparator. This amplitude comparator 

determines whether the correlation level is above or below a predetermined 

acceptable value. If the level is acceptable, the output filter amplifier is 

allowed to accept the velocity analog voltage from the phase-lock loop. If 

the correlation level is momentarily substandard for any reasons, the output 

filter amplifier is automatically thrown into memory for the duration of the 

unacceptable correlation condition. 

In operation of any type of image velocity sensor occasional short periods 

of time will occur when the image signal-to-noise ratio falls below the threshold 

required for acceptable operation. By the use of signal correlator and memory 

feature of the output filter amplifier, the HG-469B sensor design maximizes 

the accuracy of image velocity output voltage. 

3-20 

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3.5 EQUIPMENT CHARACTERISTICS 

3.5.1 Size. The Beta System consists of two major components: the 

detector and the electronics package. The detector shell approximates a 

barrel shape (See Figure 1-3) with a maximum diameter of 10 inches and a 

length, including adaptor, of 15.2 inches. The electronic package is a 

rectangular box, with optimum dimensions of 9 inches (length), 6 inches 

(width) and 7 inches (height). Several reductions in size may be made, 

with added design time, to accomplish a package of 9" x 6" x 6" as 

specified. 

3.5.2 Weight. The following table lists major subassembly weights 

for the Beta System. For,comparison purposes, the weights of the present 

configuration are listed as well as a revised configuration utilizing the 

single path herringbone concept involving simplified optics (see Appendix 

A2). The weights are realistic estimates using basic aluminum structures 

for low cost fabrication. Further refinements utilizing beryllium and/or 

magnesium structures for reduced weights with increased fabrication costs 

are in process. 

TABLE 3-1. 

Detector Present Configuration Herringbone 

Optics 12.00 4.00 

Structure/Support 7.00 6.25 

Electronics 1.00 1.50 

SUBTOTAL 

Electronic Package 

20.00 11.75 

Electronics 4.00 4.00 

Structure/Support 4.25 4.25 

Low-Voltage Power Supply 1.75 1.75 

SUBTOTAL 10.00 10.00 

HG-469B TOTAL 30.00 21.75 

-see-RH-Ifs 3-21 Page .... /9! 
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Volume II 

A/D Converter 

Present Configuration 

1.00 

Supporting Structure .25 

Added Connectors, etc. .25 

SUBTOTAL 1.50 

Herringbone. 

1.00 

.25 

.25 

'1.50 

Estimated Grand Total Weight 31.50 lbs. 23.25 lbs. 

3.5.3 Power Requirements 

The total estimated (from actual breadboard measurements) power required 

for the HG-469B Beta System is 26 watts without the A/D Converter. This 

power will remain virtually constant regardless of the input voltage, 

(within the voltage specification limits). The following tabulation 

gives a power breakdown: 

TABTR 3-2 

Detector 

Reference Light Assembly 0.28 watts 

Disc Motor 2.56 

Photomultipliers and Power Supply 2.00 

Preamplifiers 0.12 

4.96 watts 

Electronics Package 

Filtering/Limiting, Reference 0.70 watts 

Filtering/Limiting, Image 1.80 

Light Level Indication 0.14 

Driver/Memory 0.80 

3-22 

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Disc Servo Amplifier 1.00 

Compensation Amplifier 2.20 

Integrator 0.20 

Output 1.10 

Power Amplifier 2.50 

30.44 watts 

Low Voltage Power Supply 10.66 watts 

TOTAL 26.00 watts 

A/D Converter 0.90 

Power Supply Efficiency 0.90 

GRAND TOTAL 27.80 watts 

Figure 3-5 illustrates the power distribution. 

3.5.4 Hardware Description. 

3.5.4.1 Detector. The detector size is constrained by the optical 

beam splitting system necessary to generate a separate channel for X 

and Y outputs. The breadboard optical system is shown in Figure 3-6, 

to the right is a wood model of the optics less supporting structure. 

Figure 3-7 shows the chopper disc, servo motor, photomultiplier assembly, 

reference sources and photodiode assembly of the breadboard (less photo 

multiplier tubes). 

3.5.4.2 Electronics Package. Figure 1-2 is an external view of 

the Prototype Engineering Evaluation Model. The electronics package 

mounts to the cold plate on the equipment rack and contains: 

1. Two identical (K and Y channel) input-output boards. 

--S-EeffET-f13" 

3-23 

Page 1/ 
469B-2 Bly_005-112_68 

Volume II 

Y LOCK-ON IND ICAT ION 

.... 

■•.1 2 

7- 0 

= .-) .-. a_ ...c 

T 

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in . , 

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3-24 

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BIF-005-112-68 

Figure 3-6. Optical Assembly 

3-25 

Page 
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Volume II 

Figure 3-7. Disc, Motor, and Photomultiplier Assembly. 

L.• 3-26 Page s'4, ....... 
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2. Two identical (X and Y channels) correlation loop circuitry. 

3. One board containing the disc servo amplifier and low light 

comparator circuitry. 

4. One board containing the interboard and input/output connections. 

5. Low voltage power supply. 

6. The CFE A/D Converter. 

7. Five connectors for the detector (1), power (1), input/output (3). 

3.6 SPECIFICATION COMPLIANCE 

The Hycon Beta System will meet the requirements of EC-701A, Rev. 6 

except for the following: (numbers refer to paragraphs of EC-701A, Rev. 6.) 

1. 3.1.1.1.1. Insert this sentence after the fourth sentence: 

"A notch filter network at 30 Hz is permitted." See Figure 3-8. 

2. 3.1.2.4 Insert this sentence after the third sentence: "The 

IVS head shall not be required to meet the 4 inch pivot .drop 

requirement of DR 1100B, Table I (Nov. 1967)." 

3. 3.2.1.2.1.1f No change except that: 1) Maximum average power 

consumed should be 28 watts including the A/D Converter. (Hycon 

Beta will not increase power consumption with increase of input 

voltage.) 2) Maximum turn-on transient energy content shall 

not exceed 100 millijoules over and above the maximum operating 

condition. 

4. 3.2.1.2.2.1 Specification EC 1602, A/D Converter, was not 

supplied with the RFT). Based on preliminary size, weight, and 

power data, Hycon can accanmodate this item within the Beta 

electronics package. 

5. 3.2.1.2.3.2 Agreed, except that height should be 7 inches. 

If the 6 inch height is mandatory, other accommodations could 

be negotiated. 

3-27 

469B-2 

Volume II 
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Volume II 

RELEASE 1 JULY 2015 

6. 3.2.1.2.3.3 Change to: "The total weight of the IVS System 

exclusive of cabling and sensor head adaptor shall not exceed 

30 pounds." (One of the options meeting the requested weight 

specifications is summarized in Section 3.5.2 and Appendix A2 

of the Proposal. The others will be in an addendum to be 

submitted at a later date.) 

7. 3.2.1.2.5.2 The mounting configuration of the sensor head for 

testing to 4.4.3.2.A of DR 1100B remains to be defined. 

8. 4.1.3.4.2 Add to end of last sentence: ". . . except that the 

vibration curve is the lower curve of Figure 8, EC-701A, 

Rev. 6." 

9. 4.1.3.4.3.B 'Change to: "Paragraph 4.4.3.5 of DR 1100B applies 

except use higher curve of Figure 8, EC-701A, Rev. 6." 

10. 4.1.3.4.3.G DR 1100B, 4.4.3.15f says these tests do not have 

to be run if the leakage requirements of 4.4.3.11 of DR 1100B 

are met. (It is assumed that the yet-to-be-determined leakage 

requirements (4.1.3.4.3.N below) will still allow an 

exception to this test.) 

11. 4.1.3.4.3.H Same as 4.1.3.4.3.G. 

12. 4.1.3.4.3.M Change "Zone 3" to "Zone 2". 

13. 4.1.3.4.3.N Expect to meet, however, test conditions are yet 

to be determined. To establish preliminary requirements and 

to meet the intent of para. 4.4.3.15f of DR 1100B, Hycon will 

assume the leakage test is that of DR 1100B. 

3-29 

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3.7 THE PROBLEM, RISKS, AND SOLUTIONS 

3.7.1 The Problem 

The Beta System requirements call for an image velocity sensor that 

measures orthogonal velocity components with high accuracy. Ultimate 

system use calls for a reliable, compact, and lightweight device. In 

addition to velocity information, the sensor must generate an image lock-on 

signal, a readiness signal, a low-light signal, a saturation signal and 

various diagnostic signals. 

The Hycon H0-469B Beta System meets all the requirements. It is a 

null-type device which detects and indicates image motion plus-or-minus from 

zero. The sensor is applicable to a closed loop servo system for maintaining true image motion compensation. 

3.7.2 The Risks 

Much of the potential risk will be avoided in that this is not a new 

design but an extension of an original design. Hycon has been producing 

the proprietary HG-453 Heterodyne Image Motion Sensor which resulted 

from an applied research and development program directed towards image 

velocity detection and true image stabilization. The program resulted 

in a new principle of image velocity measurement based on heterodyne 

and correlation concepts. The HG-453 uses scene scanning at enhanced 

rates to overcome the dynamic limitations of previous fixed grid sensors. 

These limitations it overcomes are, limited bandwidth and the inability 

to distinguish between positive and negative velocity components. 

The HG-469B Beta System consists of essentially two V/R Sensors 

inside a single package. One half detects and indicates the forward 

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motion components, the other half detects and indicates the cross-track. 

motion. It is of critical importance that both halves of the sensor 

view the same scene at the same time and this requires a suitable beam 

splitting optical system. 

3.7.2.1 Eccentricity of the Sensor Disc. An apparent risk might be the 

degree of concentricity required for the sensor disc since an eccentricity 

would result in a second order error in the detected image motion rates. 

However, this problem has been minimized by use of: 1) a servo motor 

utilizing large,precision,external bearings for improved alignment and 

long life; 2) a highly accurate, deposited metal on glass disc pattern; 

3) a pair of reference lights on each side of the detector and at the 

same radius as the detector to balance out any lead and lag error caused 

by eccentricity. In addition, the disc is mounted on the shaft by a 

microscope positioning technique such that the residual eccentricity 

of the disc is less than 1/10,000 of an inch. 

3.7.2.2 Photomultiplier Tube. Because this item is a cold cathode vacuum 

device, the reliability is better than a filament device, however it is 

not equal to a semiconductor device. Since input light levels dictate 

use of this part, critical study and careful design must be depended on 

to reduce risks. Several vendors have supplied photomultiplier tubes 

for other programs with environmental conditions similar to that required 

for the Beta environment. In addition, packaging methods will be used 

to minimize vibration and high temperature problems associated with this 

type of device. The corona and other high voltage problems normally 

encountered will be eliminated by careful design of the package. 

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Volume II 

3.7.2.3 Optics. The risks in the optical system design lie mainly in the 

environmental areas. Extremely careful design is being accomplished to 

allow the optics to perform within specification after being subjected 

to temperature and vibration extremes. This design is also complicated 

by the need to minimize weight. Prototype efforts to date have shown 

that all objectives on this item are achievable. 

3.7.2.4 Hermetic Seal. Use ofahermetic seal on the Beta is the subject 

of a trade-off study in progress. Positive factors are minimization of: 

atmosphere compatibility problems, humidity problems, high voltage and 

corona problems, piece-part selection problems. Negative factors are: 

structure and interface design complexity to allow reliable sealing of 

the units, including a field lens seal in the sensor head. Prototype 

packaging has shown that careful design will minimize risks and extensive 

testing will be used to enhance reliability. 

3.7.2.5 Interface. A potential high risk relates to the interface of the 

sensor with the prime system. The mockup, breadboard, and prototype 

deliverable items reduce this risk. A common understanding of all facets 

of the relationship of the sensor to the prime system, including mechanical, 

optical, electrical, and thermal environment parameters, is necessary. 

The liaison program should continually update the interface information. 

3.7.2.6 Schedule. The long-lead time required for procurement of high 

reliability components and preliminary qualification and testing of special 

parts and assemblies may not be compatible with the present schedule without 

added early endeavor and agreement on interface and parts. Hycon will 

attempt in all respects to design the Beta to minimize risks associated 

with this problem. 

3-32 2.1fLir • CNi .17" e ,f2r 

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Volume II 

3.8 A/D CONVERTER 

This is specified to be a CFE item with the following 

preliminary characteristics: 

a. Size: 4 x 3 x 1.5 inches 

b. Weight: 1 pound 

c. Input Power: +12 VDC +5% 0.15 watt,+ 4 VDC +5% 0.7 watt 

d. Reference Voltage: +5 VDC +1%, 10 ma, 

e. Output Capacity: 24 bits serial consisting of 10 bits each 

channel plus 4 bits sign. 

This item will be installed into the electronics package and interfaced 

with the existing circuitry. The required power will 'be supplied from 

the existing low voltage power supply which will be upgraded 1.8 watts 

to allow for 0.9 watts to the A/D Converter and 0.9 watts for power supply efficiency and safety factor. 

Since only the +12 VDC of the input voltages presently exist, circuitry 

will be added to the present Beta to generate the required voltages. This 

will take the form of zener regulators for the +5% power and an integrated 

circuit regulator for the +1% reference voltage. Electronics package 

weight will increase 1.5 pounds, 1 pound for the A/D Converter and .5 

pound for the added structure and hardware.. 

3.9 TESTING 

3.9.1 Self-Test Capability 

3.9.1.1 Purpose. The purpose of the self-test capability is to provide 

a confidence test of Beta System operation, both in ground test and 

during a mission. A confidence test gives evidence that the system is 

in working condition, but it is not intended to serve as a calibration 

test certifying to quantitative aspects of operation. Accordingly, the 

test method need not be capable of quantitative measurement to the specified 

Poi 

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Beta System precision. Since proper functioning of the Beta System involves the photodetectors and spinning grid as well as the subsequent 

electronic circuitry, the self-test method should introduce the stimuli 

optically so that all system elements participate in the test. While it 

might be possible to image a moving test pattern through the Beta System 

optical train, the mechanization would involve mechanical drives and added 

optical elements which add to weight and size. For these reasons, we 

propose a test method which creates an illusion of pattern motion due 

to the modulation of intensity of four discrete point light sources 

which can be imaged by the existing Beta optical system. 

3.9.1.2 Method. A given fixed point source of light, when imaged upon 

the spinning grid of the Beta System, produces an electrical component 

in the output of the photomultiplier which is sinusoidal in waveform 

(after filtering), has exactly the reference frequency, and has a phase 

which corresponds to its (image) position on the grid relative to the 

position of the fixed reference image (the reference lamp). If the grid 

is thought of as being stationary with one of its lines placed exactly 

upon the reference position, which may be called the.zero-degree phase 

condition, then as the test point source is moved relative to the (stationary) grid, the test point phase will change by 360° with each traverse 

of a complete grid cycle. This is the principle of velocity measurement 

employed in the Beta System. 

By using four fixed light sources placed in quadrature (0°, 900, 180°, 

270°) with respect to the grid cycles and modulating the intensity of the 

sources in time quadrature, an illusion of a single point source can be 

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created moving across the grid at uniform velocity. The effective velocity of the point is the grid period multiplied by the frequency of modulation of the light sources. The direction of the effective motion is determined by the phasing of the modulation signals applied. The situation 

is entirely similar to the generation of a "rotating electric field" 

in induction motors by the time quadrature excitation of electrical 

windings in space quadrature. In our case, four light sources are used, 

rather than two, because the lights cannot generate negative intensities. 

Figure 3-9 illustrates the possible location of a reference source 

and a test source within the aperture and relative to the grid, and the 

resulting photosignals are shown at the left of the figure. Figure 3-10 

shows the manner in which photosignals of different phase with respect 

to the reference may be represented by directed vectors or "phases". 

Figure 3-11 depicts the generation of a resultant photosignal equivalent to a point moving at constant velocity (constant phase rate or 

phasor rotation) by the summation (at the photomultiplier) of the light 

received from four fixed light sources whose intensities are appropriately 

modulated. Figure 3-12 shows that each modulation is half-wave sinusoidal, 

obviating negative intensity requirements, and the locations of the images 

of the fixed sources, A B C and D, are such as to yield the phases (relative to one another) 0°, 900,, 180°, 2700. The sense of the motion 

(clockwise or counterclockwise) is determined by the sign chosen for the 

quadrature component of the modulation. Any one or more such fixed sources 

unmodulated in intensity represents a zero-velocity condition. 

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Figure 3-10. Phasor Representation 

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GRID (57,972aM/W)-111/AT 

Figure 3-9. Effect of Test Point Position Upon Phase of 

Generated Photosignal 
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Figure 3-11. Effective Phase Progression (Rotation) of Resultant Photosignal 

Fhasor (Shown Dashed) Due to Superposition of Photosignal Cordponents from Four Fixed Sources, Modulated in Intensity (Positive Only), in Space Quadrature. 

44,2Z7ILT/7)/ . 

Figure 3-12. Generation of Full Wave By Addition of Positive-only Waves. 

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3.9.1.3 Physical Configuration. The four test light sources are placed 

in a plane which is imaged on the spinning disc as illustrated in Figure 

3-13. The Beta System uses a beam splitting prism to form two separate 

(X and Y) channels. The fourth side of the prism (in plan) is currently 

unused (opaqued) and may be used as the input surface for the test-point 

array. Since the optical system normal images the input aperture upon 

the disc, a subsidiary field lens may be required to place the test-point 

array at a similar focus. 

The test light sources are silicon carbide visible light point sources 

which are positioned in the input plane so as to generate photosignal 

0 

responses at phases 0 , 900, 1800, and 270 (for A,.33, C, and D, respectively) in both channels. Thus one test array serves to test both channels. A fraction of the light from each test source is diverted to a 

photosensor,the output of which is used to servo the source intensity 

to the desired modulation signal. There is one such servo photosensor 

for each of the four test sources. 

The light from the test sources is split by the beam splitter to form 

two test patterns. When the test sources are not lighted, the Beta 

System will not experience any effect from the presence of the test array. 

3.9.1.4 Intensity Modulation. Since the grid period subtends about 

0.1 inch at the input aperture, a modulation frequency of 1 Hz will produce an apparent velocity of 0.1 inch per second. The modulating signals 

are derived from an oscillator composed of three operational amplifiers, 

the first two being used as integrators and the third as an inverter. 

3-38 

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The configuration is commonplace in analog computing. (See Figure 3-14.) 

If the output of the first amplifier is taken to be the cosine wave (all 

waveforms are sinusoidal and of equal amplitude and frequency), the 

second amplifier yields the negative sine and the third the positive sine. 

The radian frequency is 1/RC. 

By using cosine and negative sine as the quadrature components for 

modulation, a forward moving test source is simulated. When the negative 

sine is replaced by the positive sine, the effective motion of the point 

is reversed. A null-velocity test source is obtained by lighting any 

one of the four sources at constant intensity. 

It cannot be depended upon that the modulating signals will be converted linearly to intensity variation for most visible light sources. 

For this reason, two additional operational amplifiers are used as servo 

amplifiers to compare the modulating signals with the photosignals generated 

by each source. Photodiodes are employed which have a highly linear relation between photocurrent and incident light intensity. The light source 

current is switched by diodes to the proper member of each pair (A-C, 

B-D) by the sign of the control signal. The emitted light produces a 

proportional photocurrent through the photosensor. The operational 

amplifier servo loop gain acts to maintain the photocurrent equal to the 

input current through the input resistor from the modulation control signal. The servo action causes the source intensity to be an accurate 

replica of the controlling sinusoidal signal. 

The switches, which are here shown schematically, will be FET solid 

state switches driven by switching circuitry used to produce the required 

self-test cycle. 

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Figure 3-14. Intensity Modulation Servos and Low Frequency Oscillator 

with. Quadrature Outputs. 3-41 Pada. .......,........_...of._...-- 

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3.9.1.5 Cycle Control. The self-test cycle is initiated by a single 

input command pulse. Upon receiving the command, the test circuitry turns 

on the power to the operational amplifiers, and after a delay to allow 

the establishment of initial conditions, steps through a cycle of NULL, 

FORWARD, NULL, REVERSE, NULL and turns itself OFF. The circuit blocks 

which control the cycle are shown in Figure 3-15. 

The self-test command pulse is received by the flip-flop (1) which is 

set by the pulse. The setting of the flip-flop causes the binaries 

(8-11) to be reset, and turns on the AND gate (7) by connection (17). 

With the flip-flop set, the power is applied to the principle portion 

of the self-test equipment, that part enclosed by the dashed line (16). 

(The binary resets may be moved to the "delay" stage to avoid troubles 

due to power turn-on lags, a practical detail.) 

A delay (3) is introduced to allow the oscillator to settle at its 

desired initial conditions as well as to_ permit power supply transients 

to settle out. At the end of the delay, the initial condition gwitches. (4) 

are opened and the low frequency oscillator (5) commences cycling. At a 

given point in each cycle (preferably a zero crossing), the detector and 

pulse former (6) emits a pulse which passes through the AND gate (7) to 

step the binary counter sequence (8-11). The two states of each binary 

are available for use by the control logic (12). The control logic 

activates and deactivates the four control switches (13) which determine 

the NULL, FORWARD and REVERSE velocity modes of the test source array 

driven by the light source servos (14). 

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The test sequence is determined (for example) as follows: 

1. The binaries start in the state 0000 due to reset by the flip-flop 

(1). At the end of the first oscillator cycle (1 sec), the counter steps 

to 0001; at the end of two cycles, it steps to 0010; and so on. During 

this first 2 sec. period a NULL velocity is required. 

2. During the next two-second interval, the counter will indicate 

0010 and 0011, and a FORWARD velocity is called for (approximately 0.1 inch 

per second). 

3. The next two-second interval produces NULL velocity again, the counter 

states being 0100 and 0101. 

4. The fourth interval causes REVERSE velocity, the counter states 

being 0110 and 0111. 

5. The final interval returns to NULL velocity, with states 1000 and 

1001. 

6. At the end of the fifth interval, the counter steps to state 1010, 

and the self-test system turns itself OFF by resetting flip-flop (1) 

which controls the power switch. 

A time picture of the self-test cycle (as described; other cycles can 

be produced by simple logical changes) is as follows: 

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The logic shows that whenever B2 

(9) is in state "0", a NULL velocity 

is called for. When B2 is in state "1", the velocity is either FORWARD, 

when B

3 

 (10) is "0", or REVERSE, when B3 

is "1". The OFF condition is 

signalled when the state 1010 is reached. The execution of such logic 

is very simple and straightforward. 

3.9.1.6 Weight, Size, and Power of Self-Test. 

1. Weight. The added weight to the detector head is estimated to 

be ten ounces. 

2. Size. The lamp assembly will be approximately 4" x 1" x -.P1. The 

added control electronics will be a printed circuit board 5" x 3" x 

1". The detector housing size should not change. 

3. Power. An added three (3) watts of power, including power supply 

efficiency, will be required, during the self test period only. 

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3.9.2 Diagnostic Signals 

The purpose of these output signals is to provide information on 

Beta status to the telemetry. They will all be dc signals normalized 

to the voltage range of 0 to +5 VDC. Study of these functions will enable 

problem isolation to a relatively small quantity of circuitry or a single 

function. The list of preliminary functions to be monitored is as follows: 

a. X channel velocity output 

b. X sense 

c. Y channel velocity output 

d. Y sense 

e. Servo control voltage 

f: X channel correlation voltage 

g. Y channel correlation voltage 

h. HV power supply control voltage 

i. X reference signal 

j. Y reference signal 

k. +15 VDC power 

1. +12 and -6 VDC power 

m. +24 vdc voltage input 

n. +4 vdc current input 

o. Detector internal temperature 

P. Electronics package internal temperature 

3.9.3 Test Hardware 

3.9.3.1 Special Test Setup. The special test setup for field test of 

of the Beta System is the same equipment to be used for acceptance test. 

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Thus, data from the acceptance tests will correlate directly with field 

test data for the same system. Figure 3-16 is an artist's concept of 

the Special Test Setup. 

The test setup consists of an input section which provides simulated 

image motion to the detector and an output section which monitors and 

records system outputs and diagnostic signals. 

Basically, the input section consists of a cathode ray tube (CRT) 

image motion simulator and the optical elements to project the moving 

image into the detector. The CRT image motion simulator is comprised of 

a standard oscilloscope, a standard low frequency oscillator and necessary 

controls and interconnections. A standard do power supply provides Beta 

System power. 

The output section consists of a digital display of A/D converter 

outputs, an array of critical signal GO, NO-GO indications, a digital 

voltmeter for monitoring output and diagnostic signal dc levels, an 

oscillograph for monitoring and recording dynamic signals, and the 

necessary test points, interconnecting cables, and input/output switching. 

This is the minimum equipment required for every test of the Beta System. 

(For diagnosis, an oscilloscope and electronic counter will be required 

at the factory.) The test setup is housed in two standard modular enclosures that bolt together and mount on casters to form a test console. 

One enclosure is a rack which drawer mounts: 

a. Output test points and A/D output display. 

b. Input controls. 

c. A storage drawer for cables, manuals, and procedural support 

data. 

U. Commerical test equipment. 

e. Cooling blower and filter. 

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Figure 3-16. Beta Special Test Setup 

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The other enclosure contains the input optics (a lehs and mirror) oscilloscope, and the working surface where the system is mounted and connected 

for test. 

The Beta electronics package rests on a heat sink on the working 

surface of the input section and cables connect inputs and outputs at 

conveniently located connectors. The detector views downward into the 

optics of the image motion simulator and connects by a cable to the electronics package. The test is conducted by applying various types and 

rates of image motion (the oscilloscope trace controlled by the low 

frequency oscillator) and by rotating the detector to predetermined 

angles to simulate various direction of motion. For each input, the 

critical GO-NO-GO indications and dc outputs would be monitored and compared with the digital readout. For selected inputs, a recording of 

diagnostic signal dynamic characteristics would be made and compared to 

a master recording for the same test conditions. The power supply will 

allow various voltage levels and conditions to be supplied to the Beta 

System. 

3.9.3.2 Special Test Set Requirement. The production schedule and 

intended usage indicates the need for three Special Test Setups, one for 

acceptance testing, one for qualification testing, and one for remote 

site operational verification. Certain general purpose test equipment will 

be required and are not items included in the special test set cost. These 

items should be relay rack mounted, and are: 

1. Digital Voltmeter - Hewlett Packard 3440A with 3441A selector 

2. Oscillograph - Brush Mod. 16-2308-00-3490 

3. Power Supply - Hewlett Packard 6265A 

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4. Oscillator - Hewlett Packard 203A 

or equivalents. 

3.9.4 Optical Bench Tester. 

Figure 3-17 illustrates the first tester used. This tester lacked 

the ability to recognize certain dynamic test problems, including scalloping (a function of the haze factor and minimized by proper aperture 

shading) and pseudo velocities introduced by scene elements entering and 

leaving the aperture with changes in obliquity. As a result, using the 

Customer's Beta Tester as a guide, an optical Bench Tester has been devised 

and breadboarded. This tester has the appearance of Figure 3-18. The 

purpose of optical portion of simulation is to present the Beta System 

under test with a ray bundle having a half angle of four degrees. The 

optics consist of three f/2.5, 12 inch Aero Ektar lenses and an aperture 

stop. The first lens in the optical system produces an image at the 

focal point of the second lens. The second lens collimates the image 

rays. The collimated bundle is passed through the aperture stop. The 

location and diameter of the aperture is such as to restrict the final 

ray bundle to four degrees. The aperture also serves another function: 

the introduction of "haze" light into the system. A small lamp is situated 

in the center of the aperture. The lamp is at the focal point of the last' 

lens in the system. The haze light is presented to the Beta System as 

a collimated bundle. The last lens directs the four degree bundle to the 

Beta System and forms an image on the Beta field lens. 

Stereo track is simulated by mounting the target scene on a rate table. 

Oblique angles during stereo track are obtained by tilting the scene on 

. the rate table prior to rotation. 

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Figure 3-17. Initial Beta Testing 

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tical Ben

ch Tester 

a 

CI) 1=4 

72" OPTICAL BENCH 

CS 

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Section 4 

MAINTAINABILITY AND RELIABILITY 

4.1 MAINTAINABILITY 

The primary design criteria dictate extremely reliable gas-filled, 

sealed units that must be returned to the factory for repair. This 

system is packaged for: (1) rapid trouble isolation using test points 

carefully selected to indicate functional loop and subassembly status, 

(2) ready replacement of plug-in circuit cards and other modular subassemblies, (3) least possible calibration time by adjustment of a few 

basic circuit values. In addition, testing time is minimized by carefully analyzing parameters to identify those that must be checked to 

prove operability at acceptance and prior to each use. 

4.1.1 Test Points 

As listed in Paragraph 3.9.2, diagnostic monitoring signals are brought 

out at a connector. These signals allow trouble isolation to the functional circuit. In nearly all cases, analyzing the signals will isolate 

the trouble to a circuit card or subassembly while the units remain 

sealed. The detector assembly can be fully diagnosed by means of these 

external signals. Inside the electronics package, testing at circuit 

board connectors allows complete diagnosis of any trouble. 

4.1.2 Modular Construction 

Each printed circuit card performs one major function. Thus replacing 

• 

a card to correct one malfunction will not invalidate calibration of 

other functions. The optics assembly and the motor disc assembly are 

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mounted in a manner that will allow replacement in minimum time and without damage to structure or assemblies. 

4.1.3 Calibration and Testing 

By selecting about five basic circuit values the system is permanently 

calibrated. Subsequent testing is reduced to its simplest form by use 

of a test set that provides optical input and monitors diagnostic signals 

as well as system outputs (including A to D Converter outputs). Certain 

parameters are proved by design qualification and will not change with 

use or differ among units. Careful selection of parameters that must 

be tested will simplify the tests and reduce test setup time while 

providing high assurance of system readiness without ambiguity. 

4.1.4 Maintainability Program 

Maintainability is designed into the system. However, it must be 

proved and maintained by a comprehensive program. Refer to Volume I 

under Section 5, Plans, for a discussion of the program including 

Evaluation, and Reporting with specification basis and example forms. 

4-2 

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4.2 RELIABILITY 

4.2.1 Introduction 

Reliability Engineering is a line function reporting to the Chief 

Engineer. The group is responsible to ensure that all contract requirementS in this discipline will be met by establishing a comprehensive 

reliability program and parts selection program early in the design 

phases and by extending the programs over the contract period. The 

following paragraphs describe the Hycon reliability program. 

4.2.2 Reliability Program 

The reliability program will be implemented corresponding to the 

development and production phases of the contract. In the development 

phases, the reliability group will assume active participation in design 

reviews including performance of reliability figure of merit analysis 

and parts selection. The program will begin with'an analysis of the 

engineering model configuration and extend this analysis and parts 

selection program to the production configuration. Every effort will 

be made to identify and correct any possible sources of unreliability 

and eliminate any unreliable parts and components in preparation for 

the preproduction and production models. The production models will 

undergo formal qualification testing. The production phase will consist 

principly of adequately monitoring the performance of the production units 

to assure that the required reliability has been met. 

The following paragraphs describe the preliminary reliability program implementation in a description of the reliability tasks, and formal 

documentation. The program as described herein will conform to R-1001F, 

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Reliability Assurance Requirements for Subcontractors. A more definitive 

program will be prepared, submitted for approval, and maintained per DRL 

MSM-R-107. 

4.2.2.1 Design Reviews. The reliability engineers to the project will 

actively participate in all design reviews and will prepare design review 

check lists, reliability stress analyses, and reliability predictions for 

aides in equipment evaluation. The check lists will be prepared as comprehensive guides for an impartial evaluation of the technical data and 

will permit determination of possible design weaknesses or reliability 

inadequacies. They will also provide assurance that applicable human 

factors have been considered and appropriately resolved. The following 

paragraphs define the design review panel and design review scheduling. 

The design review panel will be composed of engineering managers, 

reliability engineers, and cognizant design engineers. The design review 

panels will consider interfaces, parts application, materials, circuits, 

processes, thermal design, mechanical design, and configuration. The 

equipment design will be carefully checked against the design review check 

list to assure that the decision reflects conformance with the basic 

specification as regards performance, reliability, and maintainability. 

When appropriate, recommendations will be made to change or approve the 

design. 

Three design reviews will be. scheduled during the design program to 

provide a periodic evaluation of each phase. Additional reviews will be 

scheduled as necessary whenever special problems are encountered. 

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a. Preliminary Design Review. A PDR will consider the detailed circuit 

analysis and layout and will be scheduled as soon as the principle design 

areas are formulated. This review will include a check of component interfaces, system block diagrams, preliminary schematics, a preliminary parts 

list, and consideration of the effects of malfunctions, failures or performance degradation of each part. 

b. Critical Design Review (CDR). The critical design review will 

emphasize design excellence and establish conformance to recommendations 

arising from the preliminary design review, and will consider any additional 

data that has become available since the preliminary design review. This 

review will be scheduled when prototype designs are available in . a packaged 

configuration. 

c. .First Article Configuration Inspection (FACT). The final design 

review will be scheduled at the end of the design phase after completion 

of the first model, of the prime hardware. Its purpose will be to review 

drawings, specifications, problem areas, manufacturing processes or procedures, overall production performance and maintenance improvement areas. 

4.2.2.2 Reliability Analysis. Reliability is an engineering design parameter. The procedures used to predict product reliability parallel those 

used to arrive at other production characteristics relating to parts or 

materials, and to the stresses anticipated in their application. A complete reliability analysis provides all calculations necessary for a 

reasonable prediction of equipment reliability and includes the use and 

application of parts, materials, and processes; a reliability figure of 

merit analysis; a failure modes and effectanalysis; and a hazard modes 

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and effects analysis, if applicable. These analyses will be prepared 

in the breadboard phase and revised as required as the design and development of the equipment progresses. The results of these analyses will be 

submitted per MSM-R-112-1 as scheduled by the DRL. 

4.2.2.3 Parts, Materials, and Processes. The selection of parts, 

materials, and processes will be controlled through the Reliability group 

for analysis and assurance that each part, material, or process selected 

will meet the requirements of the equipment specification. In the electronic circuits, microelectronics will be used wherever feasible. In 

those instances where it is not feasible, discrete components parts may 

be used. Those elements and/or parts shall be selected in accordance with 

Contractor Specification DR 1110, Selected Parts List. In the event it 

becomes necessary to use parts which are not in DR 1110, data will be 

furnished in accordance with the requirement of DR 1110 for possible inclusion as a preferred part. Where discrete parts are required, Rycon 

will obtain approval from the Contractor prior to their use. This includes 

approval of the packaging technique to be employed. All other parts, 

materials, and processes shall be selected in the following order of 

precedence: 

Contractor Specification Numbers DR1100, DR 1112, DR 1113, DR 1115 

Applicable portion of MIL-E-25366 and MIL-E-5400 

Prior Use 

Commercially available standard MIL, AN, and MS electronic and 

electrical parts. 

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All selected parts not on the Contractor preferred list (DR 1110) 

shall be submitted to the Contractor for approval per the requirenent of 

MSM-S-126-1, Request for Approval, Parts, Materials, or Processes. 

4.2.2.4 Reliability Figure of Merit Analysis. A reliability figure of 

merit analysis shall be developed following the procedures of TRA-873-74, 

dated 1 September 1961. The figure of merit analysis will include the 

development of a reliability block diagram and a mathematical model. 

Information for the models will be developed by analyzing all the 

functional parts and listing these parts with their corresponding operational stress levels. For each part application, a failure rate shall 

be stated using failure rates from MIL-HDBK-217 for electrical and electronic parts, and SP 63-470, Vol. 1-FARADA for all other parts. If other 

rates are used, they will be justified by data derived from tests of prior 

proven experience in equivalent applications. With the data derived 

above, the MTBF for the equipment will be computed and submitted as part 

of DRL MSM-R-112-1, Reliability and Safety Analysis Report. 

4.2.2.5 Failure Modes and Effects Analysis. A failure modes and effects 

analysis will be conducted on the system and documented as part of DRL 

MSM-R-112-1, Reliability and Safety Analysis Report. The failure modes 

and effects analysis will identify all failure modes of each critical 

part, material, and process and list probability of occurrence, most 

likely condition of failure, and effect on the performance of the hardware. 

4.2.2.6 Hazard Modes and Effects Analysis. A hazard modes and effects 

analysis will be conducted to result in a systematic examination of all 

the apparent ways that an equipment malfunction, material degradation, 

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improper environment, or operational misuse may generate a hazard to the 

Beta System, associated or adjacent equipment, or to the user. This report 

will include a logic block diagram, showing hardware functions, their 

interrelations and the external interfaces. Each block shall be analyzed 

to ensure a safe application of the end-item. The hazard modes and effects 

analysis will be supplied as part of DRL MSM-R-112-1, Reliability and 

Safety Analysis Report. 

4.2.2.7 Reliability Data. All failure and failure analysis reports 

resulting from development subsystem testing, acceptance, and qualification 

testing will be accumulated and summarized as a tool for corrective action 

initiation and follow-up. In addition, on each deliverable component the 

operating time will be recorded in summary reports and component logbooks 

as defined by DRL MSM-P-101. 

The following paragraphs describe the failure report and analysis system. The failure summary reports and the corrective action information 

flow functions. 

a. Failure and Failure Analysis Reports. During development testing 

of subsystem, acceptance test of the system, and qualification test programs, the failure report (Figure 4-8) and the failure analysis report 

(Figure 4-9) will be maintained. These reports will be submitted in 

accordance with DRL MSM-R-102-1. 

b. Failure Summaries. Failure summaries will be prepared at monthly 

intervals during the life of a program and will be prepared for each system 

and subsystem. Since all failures that occur are not necessarily attributable 

to the failure of a specific system or subsystem, the sifting of pertinent 

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FAILURE REPORT 

1 NEXT ASSEMBLY 5/N 2 MODEL 3 DATE 4 REPORT 

S NAME 6 CONTRACT 10 ESTIMATED OPERATING TIME 

CYCLES HOURS 

7 FAILED ITEM NAME 8 PART NUMBER 9 SYMBOL 

THIS TEST: 

ACCUMULATED . 

11 FAILED DURING: (CHECK) 

D ACCEPTANCE TEST [INSPECTION 

p OUAL I HCATION TEST E FLIGHT 

OTHER: 

12 CAUSE OF FAILURE: (CHECK) 

DASSEMBLY 0 PART [:3mATERIAL 

DWORKMANSHIP 0 DESIGN E TEST FOUIPMENT 

OTHER: 

13 SYMPTOMS AND CAUSE OF FAILURE II PRODUCTION 

STAMP 

IS INSPECTION 16 CUSTOMER 

STAMP DATE STAMP DATE 

17 EFFECT OF FAILURE 

II ACTION AND REMARKS 

19 RECOMMENOATIONS 

70 REF, TwX OR TELECOM: 

21 DISPOSITION: (CHECK) 

0 REWORK OREPLACE 05CRAP 

p hIRB [DOTHER: 0 ENGINEERiNG ANALYSIS 

22 ENVIRONMENTAL CONDITIONS 

TEMP. HUMIDITY 

ALTITUDE 

OTHER: 

23 TIME OF FAILURE 24 TEST LOCATION 23 TEST CONDUCTOR 

wuC ..11 A,g, 

Figure 4-8. 

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FAILURE ANALYSIS REPORT 

1. REPORT NUMBER 

2. FAILED PART NAME 3. SYSTEM NAME 4. CONTRACT NO. 5. FAILURE REPORT DATE 

6. SYMBOL 7. MODEL NUMBER 8. CLFST. DATA REG. NO O. PREVIOUS FAILURE RIPORTS 

10. FAILED PART NUMBER II. SYSTEM PART NUMBER 12. OPERATING USAGE 

I. CYCLES 2. HOURS 

13. FAILED PART SER. NO. 14. SYSTEM SER. NO. THIS TEST 

TOTAL ACCUM. 

15. CAUSE OF FAILURE (CHECK) 

I O ASSEMBLY 5 

2 [J PART 6 

3 17_3 MATERIAL 7 

II 

0 

WORKMANSHIP 

DESIGN 

OTHER 

16. STRESS CAUSING DAMAGE CHECK) 

I ❑ ELECTRICAL 5 

2 MECHANICAL 6 

3 ❑ PRESSURE 7 

0 HUMIDITY 

OVERLOAD 

OTHER 

4 TEST EQUIPMENT 4 II TEMPERATURE 

17. DESCRIPTION OF FAILURE(ATTACH REPRODUCIBLE PHOTOS OR DRAWINGS WHEN POSSIBLEI 

16. RECOMMENDED CORRECTIVE ACTION (INCLUDE DETAILS IN SEQUENCE OF OpERATIONI 

IS. SIGNED 20. DATE 21. CARR ECN NO. 22. 

PAGE OF 

. 5/65 

Figure 4-9. 

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failure data is necessary. The failure summaries, however, will list the 

failures that occur by system or subsystem. They will identify each failure . 

and failure report number, and describe the disposition. 

4.2.3 Preliminary Reliability Analysis 

The preliminary analysis of the failure rates that can be expected for 

the Beta System indicates that the reliability goal of 10,000 hours MTBF 

is achievable with a concentrated reliability effort and a comprehensive 

part selection program which will include minimizing operating levels and 

screening of all individual parts. 

This conclusion is based on a preliminary analysis (see Table 4-3) 

accomplished by evaluating the prototype unit. The results indicate a 

reliability of greater than 21,000 hours (MTBF) with the following 

assumptions: 

1) All parts are thoroughly screened and properly stressed to 

realize minimum failure rates for each individual part. 

2) Application factor of 1.0 has been considered in all calculations. 

3) Mechanical parts and electrical connectors were considered to 

have negligible failure rates. 

In actual practices some degradationof the calculated failure rates 

can be expected, however, these must be controlled to maintain the minimum 

reliability goal of 10,000 hours MTBF. Under a controlled reliability 

program, the 10,000 (MTBF) is an achievable goal which will be met by 

continuous application of reliability principles in the design and develop-- 

ment stages. 

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dr-7. 

TABTE 4-3 

PRELIMINARY FAILURE RATE ANALYSIS 

(Failures per Million Hours) 

PART FAILURE RITE QUANTITY. SUBTOTAL 

a ue source * 

Detector Unit 

Resistor, MIL-R-55182 .0011 A 43 .0473 

Resistor, MIL-R-26 .006o A 2 .0120 

Resistor, MIL-R-39008 .002 A 14 .0280 

Capacitor, MIL-C-39014 .005 A 2 .0100 

Capacitor, MIL-C-5 .0003 A 4 .0012 

Capacitor, MIL-C-39001 .005 A 4 .0200 

Capacitor, MIL-C-38102 .005 A 9 .0450 

Diode, TX .100 1 .1000 

Diode, Muer .600 2 1.2000 

Diode, Power Rect. .280 8 2.2400 

Photodiode .3oo 4 1.2000 

Light Source .3oo C 4 1.2000 

Transistor .100 A 2 .2000 

Micromodules .050 A 5 .2500 

Photomultiplier .625 C 2 1.2500 

Servo Motor 17/Large Bearings .5oo D 1 .5000 

Coil, MIL-T-21038 .300 A 4 1.2000 

Power, Supply, Hi-Volts 2.400 A (Parts Count) 1 2.4000 

TOT a,, DETECTOR UNIT 11.7535 

* A = MIL-HDE-4-217 

C = Manufacturer 

D = Estimated 

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TABLE 4-3 (cont.) 

PRELIMINARY FAILURE RATE ANALYSIS 

(Failures per Million Hours) 

PART FAILURE RATE QUANTITY SUBTOTAL 

Value Source* 

"X" and "Y" Channels 

Resistor, MIL-R-55182 .0011 A 358 .394 

Resistor, MIL-R-26 .0100 A 8 .080 

Resistor, MIL-R-39008 .0020 A 18 .036 

Capacitor, MIL-C-39014 .0050 A 106 .530 

Capacitor, MIL-C-39003 .002 A 10 .020 

Capacitor, Polystyrene .002 A 10 .020 

Capacitor, Ceramic .002 A 56 .112 

Diode, TX .100 A 82 8.200 

Diode, Zener .600 A 10 6.000 

Transistors, TX .100 A 48 4'.800 

Micromodules .05 A 44 2.200 

Filters, Hybrid Active .16 A 8 1.280 

Transformer, MIL-T-21038 .3 A 2 .600 

"X" and 'Y" CHANNEL TOTAL 24.272 

A = MIL-HDBK-217A 

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.SECRETP 

TABLE 4-3 (cont.) 

PRELIMINARY FAILURE RATE ANALYSIS 

(Failures per Million Hours) 

PART FAILURE RATE QUANTITY SUBTOTAL 

Value Source * 

Motor, Speed Control/ 

Pwr. Supplies 

Resistors, MIL-R-55182 .0011 A 60 .066 

Capacitors, Polystyrene .002 A 10 .020 

Capacitors, MIL-C-39014 .005 A 24 .120 

Capacitors, MIL-C-39003 .002 A 5 .010 

Diodes, TX .100 A 9 .900 

Diodes, Zener .6 A 3 1.800 

Transistors .1 A 6 .600 

Micromodules .05 A 11 .550 

Low Voltage Power Supply 6.4 A (Parts Count) 

1 

1 6.400 

ASSEMBLY FAILURE RATE 10.466 

TOTAL SYSTEM FAILURE RATE 

Detector 11.754 

"X" and "Y" Channel 24.272 

Motor Control and Power Supplies 10.466 

BETA SYSTEM FAILURE RATE 46.492 faj1ures/ 

100 hours or 

A. = MIL-HDBK-217A 

21,509 hrs. MTBF 

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APPENDIX 1 

BETA SIGNAL SLOPE LIMITER. 

The first portion of this report was submitted as an action item 

resulting from the Technical Direction Meeting of 9 - 10 May 1968. 

The second portion is an addendum resulting from Contractor direction 

at the Technical Direction Meeting of 27 June'1968. 

The concepts discussed are not a part of the Engineering Prototype 

Evaluation Model (EPEM) now under fabrication. 

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BETA SIGNAL 

SLOPE LIMITER 

REPORT ON 

TD ACTION ITEM 

1B 

(TD MEETING 9-10 MAY 1968) 

A1-2 

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It is desirable to limit the maximum rate of change of Beta output 

voltage. Since the magnitude of Beta output voltage corresponds to 

velocity, the rate of change (slope) of this voltage indicates accelera7 

tion. 

Before considering the detailed means by which acceleration limiting 

can be implemented, a brief look at the scale factors is desiiable. The 

Beta velocity scale factor is 10 volts per inch per second, or 

0.1 volts = 0.01 inches/sec. 

Assuming we choose to limit indicated acceleration to 0.001 inches/second

2 

 , 

the corresponding output voltage slope must not exceed 0.1 volts/second. 

Figure 1 shows a suitable method for limiting Beta output to any predetermined maximum acceleration. The section of Figure 1 contained within 

the dotted rectangle performs this function. The remaining structures 

depicted in the upper third of Figure 1 are existing circuitry in the Beta 

unit. Operational amplifier AR-5 is used to provide single pole law . 

pass filtering and memory functions, in addition to voltage gain. The 

voltage gain of AR-5 is established by the ratio of Rf/Ri, the pole 

location is established by the RfCf product, and the memory function is 

provided by the "gate" field-effect transistor. The proposed modification achieves its slope limiting action by providing strong negative 

feedback around AR-5 whenever the derivative of AR-5's output voltage 

exceeds a threshold value established by circuit parameters. As can 

be seen by inspection of the proposed modification, these parameters 

include the differentiator network RC product, the gain of AR-10 (as 

determined by the R2/R1 ratio), and the threshold voltage of the series 

diodes. 

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tt Qa. 

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1;1 

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The transfer function of the differentiator network is: 

ed 

ei (s) L-s (s + I/EC) 

To evaluatetheed 

 resulting from a ramp input voltage e. having a slope 

of 0.1 volts per second we write the following Laplace transformed 

relationship: 

.1 .1 

e

d (S) = —2 (S + 1/RC) S (S + 1/RC) 

• 

In order to obtain edas a function. of time we perform an inverse transformation as follows: 

1 .1 

e

d (t) = 

j:- 

S (S + 1/RC) 

Employing partial-fraction expansion: 

K

1 o 1 

S 

K

o 

(S + Y ) (S 

S=0 

1 8=-Y 

+ y ) 

1 

S (s + y) 

1 (S + 

S (S + 

)1 

Hence: 

[

-.1 RC 1 

(S + 1/RC)j 

-t/RC -t/RC 

e

d (t) .1 RC -.I RC e .1 RC (1 - e 

e

d (t) 

Al-5 

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The above result shows that a Beta output voltage slope corresponding 

to an acceleration of 0.01 in sec generates a step voltage output of t 

0.1 RC volts at the input to AR-10. Since silicon diodes D1 and D2 

have 

a threshold voltage of 0.6 volts, the gain of amplifier AR-10 should be 

adjuSted so that the step voltage input magnitude of 0.1 RC volts will 

result in an output of 0.6 volts. 

The circuit as described in Figure 1 will achieve the desired slope 

limiting for the Beta sensor signal. In the existing breadboard No. 2 

this circuit has shown its value by limiting the signal output "lumps" 

to a slope less than 0.1 volts/second. The use of the "AcquisitionTrack" switch allows the effect to be studied by placing the circuit 

in and out of use. 

Al-6 

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BETA SIGNAL SLOPE LIMITER 

WITH RAPID RETURN TO ZERO 

(ADDENDUM TO THE BETA SIGNAL 

SLOPE LIMITER PAPER OF 22 MAY 1968) 

A1-7 

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1 A previous paper has described a circuit modification which is 

effective in limiting the maximum rate of change of Beta output voltage. 

This circuit has the property of providing equal maximum slopes for 

voltages both departing from zero and returning to zero. A further 

refinement consists in modifying the limiter circuit so that the slope 

of voltages departing in either direction from zero (positive or negative going) is restrained to some predetermined moderate value, while 

the slope of recovery toward zero, either positive or negative, is 

allowed to be much greater. 

Figure 1 presents a configuration which incorporates this refinement. The portion of Figure 1 shown above the dashed line is the basic 

Beta output section. The portion of Figure 1 below the dashed line 

shows the additional circuitry required for the slope limiting function. 

Operation of the slope limiter circuit can be most easily understood 

from the viewpoint of current limiting. The current that is being 

limited is the current flowing through CD. This current, i

c

, flows 

through RDto produce voltage eD. 

Voltage eDis amplified by AR-10 whose gain is accurately established 

by the negative feedback network composed of and R2. When the field 

effect transistor switch (FET 3) is open the gain of AR-10 is: 

R1 + R2 A - R1 

1. BETA SIGNAL SLOPE LIMITER. Report on TD Action Item lB (Technical 

Direction Meeting 9-10 May 1968) dated 22 May 1968. 

A1-8 

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0 

0 

0 N H

H 

A 

H 

H H. 

p, 

0 

fa 

0 

ya 

<

0 

Re setGate 

54 

U, 

• IX ,— 

0 

U— " 

11.1 n 

cil , 

CI. L11 

CL CO 

< Lu< 

Ct LU 

Z CK 

>4 H 

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0 ..: 

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0 

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P' 0 fri 0 

.61. 

44 

(24 

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is max. = eD max. 

RD 

6o x lo-3 

- 1 ita 

60 x 103 . 

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For the values shown in Figure 1 the gain is 10. When FET 3 is closed 

the gain of AR-10 is unity. The current limiting level is established 

by the forward threshold of the silicon diodes D1 and D2, the gain of 

AR-10, and the value of RD. For silicon diodes the threshold voltage 

is approximately 600 millivolts which results in eDhaving a value of 

60 millivolts under limiting conditions. Since eDis produced by is 

flowing through the current sampling resistor RD, we can write: 

This maximum current of one microampere flowing through CDproduces a 

voltage drop across C

D 

which is: 

1 

= dt 

Substituting values we get: 

1 

ec -6 x 

D 10 x 10 

1x 10 = 0.1t 

The voltage e. is the sum of the voltages appearing across CD 

and RD, 

Hence: 

el . = e

D 

 e0 = 0.060 + o.lt 

, D 

The above result shows that under limiting conditions (for signals of 

'either polarity departing from zero and FET 3 open) the voltage e and 

the /3 output voltage will be a ramp with a slope of 0.1 volts per second 

• 

superimposed on a baseline value of 60 millivolts. This corresponds to 

an indicated maximum output acceleration of 0.01 inches per second 

squared. 

• 

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In order to provide the "rapid recovery toward zero" characteristic 

the raw velocity analog voltage produced by the phase lock loop is appropriately sampled via FET 2 and has filter network R3C3. The resultant 

gated velocity analog is sensed by the null zone detector, which is a 

bipolar amplitude comparator. Whenever the velocity analog is within a 

predetermined plus or minus region about zero, the null zone detector 

output causes the FET driver to activate FET 3 which in turn shorts out 

R2. Shorting R2 produces unity gain of AR-10. This forces er, to assume 

a value of 600 millivolts during the recovery to zero, as opposed to the 

60 millivolts existing under limiting conditions for signals departing 

from zero. During the recovery toward zero the capacitor current is 

increased tenfold relative to its value during departure from zero, hence 

the i3 output voltage recovery toward zero will be rapid. 

The described circuit provides a means for differential limiting as 

a function of departure from, or recovery toward zero. It should be 

noted that the track switch S1 provides a means for disabling the limiter 

action during acquisition periods when large signal dynamics prevail. 

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APPENDIX 2 

Special Report on the HERRINGBONE IMAGE RATE SENSOR SYSTEM. 

The following report is of special interest since this concept 

coupled with a much simpler optical subsystem (one-path) results in 

a system that has equivalent performance, yet will meet the weight 

specifications of EC-701A, Rev. 6. This weight is compared with 

the two-path version in Section 3.5.2 of this proposal. 

In preparation is a Hycon paper defining a dual-path Herringbone 

Image Rate Sensor System. Such a system promises to minimize "dropouts" as an alternate to the approach taken with slope limiting. 

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SPECIAL REPORT ON THE 

HERRINGBONE IMAGE RATE 

SENSOR SYSTEM 

Prepared Under AFE No. 9026-01 

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INTRODUCTION 

The herringbone two orthogonal axes image motion sensor is a system 

which detects two separate velocity channels using a single channel 

optical input and single photo sensor. This is accomplished by superimposing two orthogonal grid patterns of different spatial frequencies, 

each of which make a 45 degree angle to a radial line through the center 

of the disc. When the disc is turned at a fixed angular rate, each pattern yields a different carrier frequency. 

The image is focused on the grid pattern, allowing the motion of the 

image to modulate the carrier frequency of the disc (Doppler effect). 

The output of the photodetector is the sum of the two image motion 

channels. These two channels are separated using standard filtering 

Lechniques and finally fed to the Beta electronics as is done with the 

present Beta sensor. Here the image channels are demodulated with 

respect to the carrier frequencies and each resultant image wave is 

converted to a dc voltage proportional to image rate for that axis. 

The inherent advantages of such a system over the existing Beta 

System are: 

a) Elimination of the beamsplitter portion of the optics (cost, 

weight and space saving). 

b) Elimination of one photo sensor and associated electronics 

(cost, weight and space saving). 

c) Increased reliability of the sensor head due to fewer parts. 

DISC DESIGN 

The basic disc design is shown in Figure 1. The aperture is shown 

with a line passing through its center. Each line passing through the 

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pt.(' el ridifzes 

7s" r /4-r- 4.0.-2. arc k 

dey_c4as 5".(5;%44,i5 

ei;pez-Ve-ef-c le") 

Figure 1. Construction of Herringbone Grid Lines 
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center of the aperture is tangent to a circle of radius 2 

times 

the radius to the center of the aperture. A by-product of this constraint is that each line will make a 45 degree angle with a radial 

line when both lines intersect at the disc radius of the aperture center. 

On one channel there are 750 dark and 750 clear spaces. This yields 

a clear space width of approximately 0.005 inches, similar to the Beta 

System. 

Width of clear space u (Disc dia. at aperture center) 42-- 

- 

2 (Number of clear spaces) 2 

The spatial frequency of the remaining channel is selected such that 

is is not a harmonic of the first, but yet emu can be separated from 

the sum of the two. Duxing the feasibility testing of the herringbone 

breadboard, the ratio 15/19 was used. The, signals were separated with 

crosstalk measured at less than 0.1%. Since it has been found that 

scene energy is concentrated at the lower spatial frequencies, a lower 

grid wave number is used for the second channel. The second channel 

has 15/19 (750) or approximately 590 cycles per revolution. 

SIGNAL PROCESSING 

Signal processing was performed on the herringbone breadboard as 

follows. The signal from the photodetector was amplified with a double 

peaked response, one peak at each carrier frequency. The higher the Q 

of the peaks, the better will be the rejection of harmonics in the 

vicinity of the carrier frequencies. By putting a notch filter at the 

higher carrier frequency, the lower image channel can be separated from 

the sum. Similarly, by putting a notch at the lower frequency, the 

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higher image channel can be selected. Achievable attenuations of active 

notch filters are in the range of 500 to 1000. 

RESULTS OF THE HERRINGBONE BREADBOARD 

The herringbone theory was reduced to practice using two radial discs 

mounted with their aperture grid lines at right angles at the detector 

field of view. The discs were geared together by the ratio 15/19. A 

photo source and a light sensitive diode were used to sense the reference 

frequency of each disc. In this manner each image rate could be measured 

with respect to its own reference frequency. As mentioned above, the 

output of the single photodetector was amplified by a double tuned 

amplifier. The channels were separated using two active notch filters. 

The results of the test were conclusive in that it is possible and 

feasible to build two patterns on one disc which are orthogOnal and capable of modulating a focused scene at a given carrier frequency. The 

quality of the output image signals suffered slightly because: 

a) The discs were geared, giving rise to short term disc speed 

variations. 

b) The input scene brightness to the photodetectOr was low because 

of the long focal length lens used on the breadboard. 

These problems notwithstanding, the breadboard sensed and separated 

the two image rates in good stead; no design or feasibility problems 

were encountered. 

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