# Patent

## Title
Methods, systems, and computer program products for parallel correlation and applications thereof

## Abstract

A fast correlator transform (FCT) algorithm and methods and systems for implementing same, correlate an encoded data word (X0-XM−1) with encoding coefficients (C0-CM−1), wherein each of (X0-XM−1) is represented by one or more bits and each said coefficient is represented by one or more bits, wherein each coefficient has k possible states, and wherein M is greater than 1. X0 is multiplied by each state (C0(0) through C0(k−1)) of the coefficient C0, thereby generating results X0C0(0) through X0C0(k−1). This is repeated for data bits (X1-XM−1) and corresponding coefficients (C1-CM−1), respectively. The results are grouped into N groups. Members of each of the N groups are added to one another, thereby generating a first layer of correlation results. The first layer of results is grouped and the members of each group are summed with one another to generate a second layer of results. This process is repeated until a final layer of results is generated. The final layer of results includes a separate correlation output for each possible state of the complete set of coefficients (C0-CM−1). The final layer of results is compared to identify a most likely code encoded on the data word. The summations can be optimized to exclude summations that would result in invalid combinations of the encoding coefficients (C0-CM−1). Substantially the same hardware can be utilized for processing in-phase and quadrature phase components of the data word (X0-XM−1). The coefficients (C0-CM−1) can represent real numbers and/or complex numbers. The coefficients (C0-CM−1) can be represented with a single bit or with multiple bits (e.g., magnitude). The coefficients (C0-CM−1) represent, for example, a cyclic code keying (“CCK”) code set substantially in accordance with IEEE 802.11 WLAN standard.

## Background
CROSS-REFERENCE TO RELATED APPLICATIONS
This patent application is a continuation-in-part of U.S. non-provisional application Ser. No. 09/987,193, titled, “Method and Apparatus for a Parallel Correlator and Applications Thereof,” filed Nov. 13, 2001, now U.S. Pat. No. 7,010,559, which claims priority to U.S. provisional application Ser. No. 60/248,001, titled, “Method and Apparatus for a Parallel Correlator and Applications Thereof,” filed Nov. 14, 2000, both of which are incorporated herein by reference in their entireties.
The following application of common assignee is related to the present application, and is herein incorporated by reference in its entirety: U.S. non-provisional application, “Method and System for Down-Converting an Electromagnetic Signal, Transforms for Same, and Aperture Relationships,” Ser. No. 09/550,644, filed Apr. 14, 2000.


BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to matched filters and correlators and, more particularly, to a novel fast correlator transform (“FCT”) and to methods and systems for implementing same.
2. Related Art
Matched Filter Theory was introduced by D. O. North in 1943. Others such as Van Vleck, Middleton, Weiner, and Turin, have added to the body of knowledge and application, ranging from RADAR to SONAR and communications. Although the basic theory was originally applied to continuous time systems and linear networks, the theory has been augmented to absorb discrete sampled filters and correlator operations. Indeed Paul Green pointed out in the June 1960 IRE Transactions on Information Theory, Matched Filter Issue:

    
    
        “Correlation detection was studied at first as a separate subject, but the equivalence of the two operations (meaning matched filters and correlators) was soon appreciated.”
    
    


IRE Transactions on Information Theory, New York, N.Y.: Professional Group on Information, Institute of Radio Engineers, June, 1960, incorporated herein by reference in its entirety.
More recently Van Whalen and Blahut as well as others have provided proofs of mathematical equivalence of correlation or matched filtering based on a change of variables formulation.
With the surge of DSP and proliferation of VLSI CMOS circuits as well as the universal push for software radios, many discrete matched filter realizations have evolved. The most notable digital implementation is the Transversal or Finite Impulse Response Filter which may realize the time flipped impulse response of the waveform to be processed or utilized as a correlator, both which produce the equivalent result at the optimum observation instant.
A particular concern arises when multiple filtering operations are required, concurrently, as is the case for parallel correlators. The complexity, clock speeds and signal flow control typically increase cost, size, and power.
What are needed are improved methods and systems for performing matched filtering and/or correlating functions, including concurrent and/or parallel correlations.
SUMMARY OF THE INVENTION
The present invention is directed to methods and systems for performing matched filtering and/or correlating functions, including concurrent and/or parallel correlations. More particularly, the present invention is directed to a fast correlator transform (FCT) algorithm and methods and systems for implementing same. The invention is useful for, among other things, correlating an encoded data word (X0-XM−1) with encoding coefficients (C0-CM−1), wherein each of (X0-XM−1) is represented by one or more bits and each coefficient is represented by one or more bits, wherein each coefficient has k possible states, and wherein M is greater than 1.
In accordance with the invention, X0 is multiplied by each state (C0(0) through C0(K−1)) of the coefficient C0, thereby generating results X0C0(0) through X0C0(K−1). This is repeated for data bits (X1-XM−1) and corresponding coefficients (C1-CM−1), respectively. The results are grouped into N groups. Combinations within each of said N groups are added to one another, thereby generating a first layer of correlation results.
The first layer of results its grouped and the members of each group are summed with one another to generate a second layer of results. This process is repeated as necessary until a final layer of results is generated. The final layer of results includes a separate correlation output for each possible state of the complete set of coefficients (C0-CM−1). The results in the final layer are compared with one another to identify the most likely encoded data word.
In an embodiment, the summations are pruned to exclude summations that would result in invalid combinations of the encoding coefficients (C0-CM−1). In an embodiment, substantially the same hardware is utilized for processing in-phase and quadrature phase components of the data word (X0-XM−1). In an embodiment, the coefficients (C0-CM−1) represent real numbers. In an alternative embodiment, the coefficients (C0-CM−1) represent complex numbers. In an embodiment, the coefficients (C0-CM−1) are represented with a single bit. Alternatively, the coefficients (C0-CM−1) are represented with multiple bits (e.g., magnitude). In an embodiment, the coefficients (C0-CM−1) represent a cyclic code keying (“CCK”) code set substantially in accordance with IEEE 802.11 WLAN standard.
Further features and advantages of the invention, as well as the structure and operation of various embodiments of the invention, are described in detail below with reference to the accompanying drawings. It is noted that the invention is not limited to the specific embodiments described herein. Such embodiments are presented herein for illustrative purposes only. Additional embodiments will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein.



BRIEF DESCRIPTION OF THE DRAWINGS/FIGURES
The present invention will be described with reference to the accompanying drawings. The drawing in which an element first appears is typically indicated by the leftmost digit(s) in the corresponding reference number.
FIG. 1 is a block diagram of an example discrete transversal matched filter or correlator, in which the present invention can be implemented.
FIG. 2A is an expanded view of a summation function 102 illustrated in FIG. 1.
FIG. 2B is a block diagram of a FCT kernel implemented in an 802.11 demodulator 210 in accordance with an aspect of the present invention.
FIG. 3A illustrates example correlation kernels, in accordance with an aspect of the present invention.
FIG. 3B illustrates example first and second layers of an FCT processing hierarchy in accordance with an aspect of the invention.
FIG. 3C illustrates an example signal flow diagram for the first and second layers illustrated in FIG. 3B.
FIG. 4A illustrates a conventional parallel correlator approach.
FIG. 4B illustrates an example matrix form of coefficients for a parallel correlator in accordance with an aspect of the present invention.
FIG. 5 illustrates an example complex fast Hadamard Transform.
FIG. 6 illustrates an example parallel magnitude compare operation in accordance with an aspect of the invention.
FIG. 7 illustrates an example final layer of an FCT processing hierarchy in accordance with an aspect of the invention.
FIG. 8 illustrates an example process flowchart for implementing an FCT algorithm in accordance with an aspect of the present invention.
FIG. 9 is a signal path diagram for an example CCK decoder output trellis, including an I signal path 902 and a Q signal path 904.
FIG. 10 is a block diagram of an inverted channel detection circuit 1000 that can be used to determine if one of the channels was inverted.
FIG. 11 is a block diagram of a complex correlation in accordance with equation 12.
FIG. 12 is a block diagram of an example FCT 1200 in accordance with the present invention.
FIG. 13 is a block diagram of another example FCT 1300 in accordance with the present invention, including stages 1a, 1b, 2a, 2b, and 3.
FIG. 14 is an example block diagram of stage 1a of the FCT 1300.
FIG. 15 is an example block diagram of stage 1b of the FCT 1300.
FIG. 16 is an example block diagram of stage 2a of the FCT 1300.
FIG. 17 is an example block diagram of stage 2b of the FCT 1300.
FIG. 18 is a block diagram of an example system 1800 that calculates An values.
FIG. 19 is a block diagram of connection stages for an example simplified architecture FCT 1900, including stages 1a, 1b, 2a, 2c, and 3.
FIG. 20 is an example block diagram of stage 1a of the FCT 1900.
FIG. 21 is an example block diagram of stage 1b of the FCT 1900.
FIG. 22 is an example block diagram of stage 2a of the FCT 1900.
FIG. 23 is an example block diagram of stage 2b of the FCT 1900.
FIGS. 24A and B are an example block diagram of stage 3 of the FCT 1900.
FIG. 25 illustrates an example structure 2500 for implementing equation 17.
FIG. 26 is another example block diagram of stages of the FCT 1900.
FIG. 27 is another example block diagram of stages of the FCT 1900.



DETAILED DESCRIPTION OF THE INVENTION
Table of Contents


    I. Introduction
    II. Example Environment: 802.11
    III. Fast Correlator Transform and Correlator Kernels
    IV. Mathematical Formulation
    V. Comparisons to the Hadamard Transform
    VI. Maximum Likelihood Decoding (AWGN, no Multipath)
    
        A. Magnitude Comparator
    
    
    VII. Example Methods for Implementing the FCT Algorithm
    VIII. CCK Chip Code Words
    IX. CCK Decoder
    
        A. Introduction
    
    
    X. CCK Correlator Optimizations
    
        A. Introduction
        B. Initial Complex Correlation
        C. Fast Correlator Transform (FCT)
        D. FCT Based on CCK Code Properties
    
    
    XI. Conclusion

I. Introduction



FIG. 1 is a block diagram of an example discrete transversal matched filter or correlator 100, also referred to herein as a finite impulse response (“FIR”) filter 100. The FIR filter 100 includes a set of multipliers 104 that multiply data words Xi by coefficients C. The FIR filter 100 also includes a final accumulate or summation function 102. The matched filter 100 implements the following discrete sampling equation;




  
    
      
        
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In Eq. 1, Xi are typically derived from a sampling device such as an A/D converter and can be represented as signed magnitude, two's complement, or other binary representation in parallel bit format. Likewise the multiplier tap coefficients C0 . . . Cn can be 1 bit values or k bit (soft) values depending on the application. In the case of soft value implementations, the multiplier functions can represent considerable hardware or algorithmic complexity. The final summation can be implemented using discrete adder technologies or floating point processors. The entire architecture can be implemented in hardware, software, or a hybrid of each.
A particular concern arises when multiple filtering operations are required, concurrently, as is the case for parallel correlators. The complexity, clock speeds and signal flow control typically increase cost, size, and power. Hence, efficient architectures are often pursued to streamline the implementation and thereby differentiate product offerings.
The present invention is directed to a novel fast correlator transform (“FCT”) algorithm that reduces the number of additions for parallel correlation, as compared to conventional techniques. The present invention is also directed to methods and systems for implementing the novel FCT algorithm. Conventional techniques are typically of less universal application and restricted to certain signaling schemes. The present invention, on the other hand, is applicable to a variety of parallel matched filter and/or correlator operations.
The present invention is as efficient or more efficient than the Fast Walsh Transform, currently applied as the “state of the art,” and is more universally applicable to signaling schemes employing higher order signaling spaces such as MQAM, CDMA, etc. In addition, classical multi-path processor algorithms are more easily applied using the classical correlator/matched filter kernel, in accordance with the present invention, rather than the conventional modified Fast Walsh Transform.
The present invention can be implemented in hardware, software, firmware, and/or combinations thereof. For example, and without limitation, the invention, or portions thereof, can be implemented in hardware logic. The invention, or portions thereof, can also be implemented in software imbedded in a computer readable medium (i.e., a computer program product), which, when executed on a computer system, causes the computer system to perform in accordance with the invention.
II. Example Environment: 802.11
The present invention is described herein as implemented in an example environment of an IEEE 802.11b 11 MBPS physical layer signaling scheme. IEEE 802.11 is a well-known communications standard and is described in, for example, “Medium Access Control (MAC) and Physical (PHY) Specifications,” ANS/IEE Std 802.11, published by IEEE, (1999Ed.), and incorporated herein by reference in its entirety.
The present invention is not, however, limited to the IEEE 802.11 communications standard. Based on the description herein, one skilled in the relevant art(s) will understand that the present invention can be implemented for a variety of other applications as well. Such other applications are within the spirit and scope of the present invention.
A distinction is made herein between a basic correlator kernel and a full demodulator. A general form of a demodulator for IEEE 802.11 typically requires four correlator banks operating essentially in parallel. A fast correlator transform (“FCT”) kernal, in accordance with the invention, typically includes similar structure, plus complex additions and subtractions from two in-phase and two quadrature-phase banks to accomplish the demodulation for IEEE 802.11. The Walsh transform, as a comparison, accounts for these additional adds and subtracts by having them built into its algorithm.
The 802.11 signaling scheme of interest is based on Cyclic Code Keying (“CCK”) derived from Walsh/Hadamard functions. A restricted set within the available coding space was selected so that the Fast Walsh Transform could be utilized to implement an efficient correlator architecture. Originally, both Harris and Lucent could not figure out how to apply a classical parallel matched filter or correlator, efficiently, to process the waveforms. The current coding space provides for 64 code words. Harris erroneously predicted that a classical parallel matched filter approach would require 8×64−512 complex additions since each code word is 8 bits, on I and Q and there are 64 code words. However, the true estimate is 7×64=448 complex additions as illustrated in the example 8-way adder tree illustrated in FIG. 2A.
FIG. 2A is an expanded view of the final accumulate or summation function 102 in FIG. 1, part of the FIR filter 100. Notice that only 7 adders are required for the example implementation. Nevertheless, 448 complex additions represent a significant number of operations. Lucent, Harris/Intersil, and Alantro apply the Fast Walsh Transform (“FWT”) to the CCK code set to reduce the correlation operation down to 112 complex multiplies due to the restriction placed on the code set.
The FWT is actually more of a decoder than a correlator. It reduces to the performance of a correlator for this specific application if the coefficients in the butterfly branches are weighted with simple hard coded values, i.e., 1, −1, j, −j. The imaginary numbers are included for the case of complex signaling.
The FCT algorithm, according to the present invention, is truly a correlation or matched filter operation and can be applied with soft value weighting or hard value weighting. Furthermore, the present invention is intuitively satisfying, possessing direct correspondence of matched filter tap weights or coefficients maintained throughout the hierarchical structure. This permits easy extension of the matched filter concept to accommodate channel equalization, MLSE, and other adaptive applications.
FIG. 2B is a block diagram of an example FCT kernel implemented in an 802.11 demodulator 210. The 802.11 demodulator 210 includes a bank of FCT kernels 212. The demodulator 210 can be reduced to 110 complex operations, with 78 of the complex operations allocated to the bank of FCT kernels 212. Additional simplification can be obtained by exploiting redundancy in the algorithm, as described below.
III. Fast Correlator Transform and Correlator Kernels
The available coding space for a 16 bit word is 216=65536. CCK whittles this space down to a code set formed by 8 in-phase (“I”) chip and 8 quadrature phase (“Q”) chip complex symbols. 64 code words are retained, carrying 6 bits of information. 2 more bits are contained in the complex QPSK representation, for a total of 8 bits of information.
Suppose then that 8 samples of an input vector X0, X1, . . . X7 are associated with optimal sampling instants from the output of a chip matched filter. Each of these samples would normally be weighted by the coefficients C0 . . . C7 then assimilated as illustrated in FIGS. 1 and 2A.
In the 802.11 example, C0 . . . C7 correspond to samples of the CCK code set symbols. The unknowns X0 . . . X7 are noisy input samples from which a specific code must be extracted. Conceptually, 64 such complex filtering operations should be performed concurrently since a specific code word cannot be predicted a priori. The largest output from the parallel bank of correlators would then be selected as the most likely code word correlation match at that particularly symbol sample time.
In accordance with the present invention, a general philosophy for correlation is imposed for partitioned segments of code words. In an embodiment, the code word correlations are divided into sub-sets. In the illustrated example embodiment, the code word correlations are divided into sample pairs. The present invention is not, however, limited to this example embodiment. In other embodiments, the code word correlations are divided into triplets, quintuplets, and/or any other suitable sub-sets.
Combinations of the code word correlation sub-sets are then provided to correlation kernels. FIG. 3A illustrates example correlation kernels 302a-302d, in accordance with an aspect of the present invention.
The correlation kernels 302a-302d represent all or substantially all possible correlation options for the first 2 samples X0, X1. In a similar fashion the remaining groupings of double samples (X2, X3), (X4, X5), (X6, X7) will also spawn their own set of 4 correlation kernels.
The number of separate correlation kernels spawned is determined by the number of correlated samples per summation, the number of correlation variations possible, and, in an embodiment, the number of invalid combinations of correlated samples. In the present example, each coefficient has two, possible states (i.e., hard value weighting). Thus each subset correlation generates 4 outputs. In alternative embodiments, soft weighting is implemented, where each coefficient is represented with multiple bits (e.g., magnitude representation).
In an embodiment, the number of correlation operations associated with binary antipodal signaling in accordance with the present invention is implemented in accordance with Eq. 2.




  
    
      
        
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The result for the example environment described above, using 2 input summers, is shown in Eq. 3:




  
    
      
        
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wherein:

    
    
        n is the number of uniquely available summer inputs;
        r is the number of summing inputs per kernel; and
        L is the number of invalid combinations.
    
    


Nk is thus the number of correlation kernels and therefore the number of adders or summers required. Groupings of two correlation samples provide for convenient binary expansion. As stated above, however, the present invention is not limited to groupings of two.
The term L represents the number of invalid or disallowed combinations. For instance, X0C0 and −X0C0 is an invalid combination when added and therefore can be subtracted from the total number of combinations. In an embodiment, 3 way adders or summers are utilized. In other embodiments, other algorithmic partitioning is utilized. For the current example, partitioning in powers of 2 is convenient and attractive in terms of potential hardware implementation.
FIG. 3B illustrates example first and second layers 304 and 306, respectively, of an FCT processing hierarchy in accordance with an aspect of the invention.
The second layer 306 includes 32 additions 308, of which 308a through 308p are illustrated, to process combinations of correlations from the first layer 304 of correlation kernels. The first layer 304 includes 16 additions related to the first 4 sample correlations X0C0 . . . X3C3, and 16 additions related to the 2nd4 sample correlations. Hence, the first layer 304 of kernels includes 16 adders 302 and the second layer possesses 32 adders 308. Once the second layer 306 is accomplished, each term that results includes 4 correlation components.
Note that 4 unique samples X0 . . . X3 at the system input spawns 24 unique 4-tuple correlations at the second layer 306 of kernel processing. The corresponding number of additions is calculated for the 4 sample correlation sequences from Eq. 4:




  
    
      
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At this point it is important to realize that all that is required is one more level of processing to obtain correlation terms consisting of 8 components which represents a full length correlation. However, it must also be recognized that there are 16 upper 4-tuple correlations as well as 16 lower 4-tuple correlations, which if exploited for all combinations in this case would yield 256 addition operations! Fortunately the CCK code set is well defined and possesses only 64 valid 8 chip component correlations. Hence, the final layer, illustrated in FIG. 7, is pruned to perform only 64 unique addition operations. Thus, a total (upper bound) number of adders used for the algorithm is:

16 (first hierarchical layer)+32 (second layer)+64 (third layer)=112

This is a remarkable result because a conventional parallel matched filter or correlator bank would require 448 complex additions. Theoretically, 112 is the upper bound. However, in practice, the Trellis may be pruned to a maximum of 78 additions on the I and 78 and the Q.
FIG. 3C illustrates an example signal flow diagram for the FCT algorithm through the first 2 layers 304 and 306. In accordance with the example above, there are 8 input samples and 32 output correlation options through the first 2 layers 304 and 306. Correlation combinations from the upper and lower 16 4-tuples provide a final trellis with 64 8-tuple options, each representing a different CCK symbol. The option having the highest output is selected during each symbol cycle as the most likely CCK symbol. In the example embodiment, the correlation operation utilizes I and Q matched filters since the symbols are complex.
IV. Mathematical Formulation
In an embodiment, a receiver in accordance with the present invention receives a waveform and an analog-to-digital converter function samples the received waveform and converts the waveform down to baseband. The received sampled waveform may be represented from Eq. 5:

Xi=Si+Ni   (Eq. 5)

Where Si represents samples from the received signal and Ni represent noise sampled from an average white Gausian noise (“AWGN”) process. This equation does not account for multi-path. The samples can be considered as complex, with I and Q components. The receiver output can be represented by Eq. 6:




  
    
      
        
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The coefficients, Ck, are considered constant for the nominal case of an AWGN channel. “n” is the FIR filter or correlator depth. For a case of m correlators operating on Xi in parallel, Eq. 6 becomes Eq. 7:




  
    
      
        
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The mth correlator branch then contains correlator coefficients uniquely related to that branch.
FIG. 4A illustrates a conventional parallel correlator approach and relates it to Eq. 7. The present invention breaks the sum over n-1 into smaller sums, typically, although not necessarily, sums of 2. The present invention applies all, or substantially all potential cross correlations and carries the 4 results forward to a subsequent level of processing. An example mathematical formulation of this cooperation is provided in Eq. 8;




  
    
      
        
          
            
              
                
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Where l,v,p,and u may be considered as indices to select different coefficients. All indices should be considered in terms of a final valid code word correlation. In the 802.11 case, 256 correlation sequences are defined by Eq. 8, but the options are sifted to correspond only to valid CCK code words. FIG. 4B illustrates an example matrix form of coefficients for a parallel correlator according to the present invention.
The coefficients all take on the values of +/−1 for the examples herein. The indices are permitted to contain zeros for consistency with the original FIR formulation. The FCT sub-matrices however are simply;




  
    
      
        
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The indices l,v,p,u are manipulated specifically to match the coefficients to the desired code words at the Y(i)l,v,p,u outputs. The indices also uniquely specify the trajectory through the signal flow path. Breaking up the original parallel matched filter coefficient matrix into the smaller 2×2 matrix permits the FCT algorithm to exploit redundant correlation operations.
V. Comparisons to the Hadamard Transform
An FCT algorithm trellis, in accordance with the present invention, is described above. A corresponding length 4 complex fast Hadamard Transform is illustrated in FIG. 5. As with the FCT algorithm, there are two such trellis structures corresponding to 8 input samples and 32 outputs. The 32 outputs include two 16 wide 4-tuple groupings, which are then utilized in combinations to produce the final 54 8-tuple correlations.
There are distinct differences between the two algorithms. For example, the FCT algorithm can function on arbitrary correlation words, even in matched filter scenarios, while the FWT requires certain signal coding structure. Also, the FCT algorithm permits greater efficiency because the number of adds may be tailored for a specific code set or application.
Harris and Lucent advertise an efficiency of 112 complex additions for CCK demodulation, which amounts to 2×112 additions. The bounding case for 64 arbitrary correlations with the FCT was shown to be 112, maximum. It turns out that the specific case of CCK may be accommodated using a pruned FCT algorithm having 78 additions on the in-phase correlation band and 78 additions on the quadrature correlation bank. An additional 36 add/subtract operations are required to complete the demodulator, for a total of 114 complex operations, versus 112 for the FWT. Additional reductions can be obtained to reduce the demodulator to approximately 100 complex add/subtract operations, as described below.
VI. Maximum Likelihood Decoding (AWGN, no Multipath)
The modified Fast Walsh/Hadamard Transform implements a complex trellis decoder for which maximum scores may be compared at the trellis output. Thus there are 64 distinct outputs of the trellis which are preferred based on Euclidean distance calculations along the trellis trajectories. Only certain specific trajectories are considered up through the second tier of the trellis. The distance properties for the decoding trellis are also Euclidean for the in-phase and quadrature phase correlators. However, it is important to realize that the total distance should be calculated within the complex plane rather than simply on I independently of Q. That is, scoring is based on Eq. 10.

Distance≡√{square root over (Iscore2+Qscore2)}  (Eq. 10)

This comes from the fact that there are pairs of I and Q chip code words which are dependent. That is the nature of the complex Walsh-Hadamard codes. Fortunately, a sufficient statistic exists which does not require the square root operation. Simply calculating the sum of the squares or estimating the vector magnitude will suffice. In this manner then the total distance or weighting through complex space is calculated. The largest output out of the 64 complex operations (weighting scenario) then becomes the most likely 8 chip complex code.
A. Magnitude Comparator
In order to determine which code symbol was most likely encoded, a magnitude compare operation is performed on the outputs from the summer 102 (FIG. 1). A variety of types of magnitude compare operations can be performed.
FIG. 6 illustrates an example parallel magnitude compare operation in accordance with an aspect of the invention. In operation, the I and Q inputs, 8 bits wide each, for example, are squared and summed at the correlator outputs to form 64 scores. These 64 scores are compared and the largest result selected as the maximum likelihood symbol estimate. Each of the 64 outputs are assigned their corresponding chip code-to-6-bit data map. An additional di-bit is decoded from the differential phase decoder. In an embodiment, the squaring operation results in 16 bit output value when the inputs from each I and Q correlator are truncated to an extent reasonable to minimize the compare tree. In an embodiment, a parallel compare tree utilizes log2(64)−1 compares to obtain the most likely result.
In an embodiment, the magnitude compare operation illustrated in FIG. 6 utilizes a flag at each level of compare to indicate the winning local score at that level. The winning local score can be traced from the output back to one of the 64 original input correlation scores to decide which 6-bit word is most likely. In an embodiment, outcomes of tie scores at one or more levels are arbitrarily determined. In an embodiment, magnitude compare operations are performed with an adder/subtractor to create the result C=A−B, where A and B are inputs.
Another magnitude compare technique that can be utilized is referred to herein as space slicing, which includes the steps of examining the MSB of the correlator outputs, and throwing away results not active in the MSB. If none are active in the MSB then the next MSB is compared, so on and so forth. Any surviving correlator outputs are compared in the next most significant MSB in succession until all necessary compares are exhausted. This technique is useful because it requires only 1-bit compares at each level down to the final compare. In an embodiment, 1 bit compares are performed with an exclusive OR gate. Generally, there is no deterministic way to predict the number of surviving compares which may be passed on to the next level, but the maximum number typically reduces by a factor of 2 at each level. This approach relies on a statistical distribution of scores, which may permit rapid sifting. If all of the distances are similar in magnitude then sifting typically requires more operations. For instance, if all 64 distance calculations/scores possess an active MSB then the first round of sifting will not eliminate any scores and all scores are then be compared in the next MSB. Although this is not likely to occur, it should be anticipated for associated hardware realization.
VII. Example Methods for Implementing the FCT Algorithm
FIG. 8 illustrates an example process flowchart 800 for implementing the FCT algorithm in accordance with an aspect of the present invention. For illustrative purposes, the flowchart 800 is described herein with reference to one or more of the drawing figures described above. The invention is not, however, limited to the examples illustrated in the drawings. Based on the description herein, one skilled in the relevant art(s) will understand that the invention can be implemented in a variety of ways.
The example process flowchart 800 illustrates a method for correlating an encoded data word (X0-XM−1) with encoding coefficients (C0-CM−1), wherein each of (X0-XM−1) is represented by one or more bits and each said coefficient is represented by one or more bits, wherein each coefficient has k possible states, wherein M is greater than 1.
The process begins with step 802, which includes multiplying X0 with each state (C0(0) through C0(K−1)) of the coefficient C0, thereby generating results X0C0(0) through X0C0(K−1). This is illustrated, for example, in FIGS. 3A, 3B, and 3C just prior to the kernels 302A, B, C, and D.
Step 804 includes repeating step 802 for data bits (X1-XM−1) and corresponding coefficients (C1-CM−1), respectively. This is illustrated, for example, in FIGS. 3B, and 3C just prior to the kernels 302E through 302Q.
Step 806 includes grouping the results of steps 802 and 804 into N groups and summing combinations within each of said N groups, thereby generating a first layer of correlation results. This is illustrated, for example, in FIGS. 3A, 3B, and 3C by the kernels 302, and the resultant first layer of results 307.
Step 808 includes grouping the results of step 806 and summing combinations of results within each group to generate one or more additional layers of results, and repeating this process until a final layer of results includes a separate correlation output for each possible state of the complete set of coefficients (C0-CM−1). This is illustrated in FIG. 3C and FIG. 7, where the summers 306 generate a second layer 310, the FCT final output trellis 702 (FIG. 7) provides separate outputs for each possible state of the complete set of coefficients (C0-CM−1) in a final layer 704.
In an embodiment, steps 806 and 808 include the step of omitting summations that would result in invalid combinations of the encoding coefficients (C0-CM−1). This is illustrated in steps 806A and 808A. This also is illustrated, for example, in FIG. 7, wherein the second level of results 310 omits the following combinations:

    
    
        C4C5C6(−C7);
        C4C5(−C6)(−C7);
        (−C4)C5C6(−C7);
        (−C4)C5(−C6)(−C7);
        C4(−C5)C6(−C7);
        C4(−C5)(−C6)(−C7);
        (−C4)(−C5)C6(−C7); and
        (−C4)(−C5)(−C6)(−C7).
    
    


In this example, the omissions eliminate performing summations for combinations that are invalid in light of the CCK code or that result in null operation. In other embodiments, different combinations may or may not be omitted based on particular codes.
Step 810 includes comparing magnitudes of said separate correlation outputs, thereby identifying a most likely code encoded on said data word. This is illustrated, for example, in FIG. 6, by the example parallel magnitude compare operation.
In an embodiment, the process flowchart 800 further includes the step of performing steps (1) through (5) using substantially the same hardware for in-phase and quadrature phase components of the data word (X0-XM−1).
In an embodiment, the coefficients (C0-CM−1) represent real numbers. In an alternative embodiment, the coefficients (C0-CM−1) represent complex numbers.
In an embodiment, the coefficients (C0-CM−1) are represented with a single bit. Alternatively, the coefficients (C0-CM−1) are represented with multiple bits (e.g., magnitude).
In an embodiment, the coefficients (C0-CM−1) represent a cyclic code keying (“CCK”) code set substantially in accordance with IEEE 802.11 WLAN standard, illustrated in the tables below.
In an embodiment, as illustrated in one or more of the prior drawing figures, M equals 8, N equal 4, and the coefficients (C0-CM−1) have two states, plus and minus.
VIII. CCK Chip Code Words
Tables are provided below that illustrate source-input data symbols, 8-bits long (d0-d7), and corresponding in-phase and quadrature phase 8 chip symbols used for CCK. The complex chip notation is provided for reference. In addition the algorithm flow diagram 4-tuple sums are provided since the last level of flow diagram becomes complex and difficult to follow. B0 . . . B31 are the 4-tuple intermediate correlation results relating to the signal flow diagrams presented for the correlator algorithm. Two branch 4-tuples form the final output result for each correlator. B0 . . . B15 provide options for the first branch component to form a final output correlator 8-tuple while B16 . . . B31 provide the second branch component or second 4-tuple. For instance, Table 1 illustrates an example build-up:








TABLE 1





4-tuple Designator
4-tuple Coefficient Sequence







B6
−1, 1, 1, −1


B28
1, 1, −1, −1


4-tuple Combination
Final 8-tuple Correlator Coefficient Sequence


B6 + B28  
−1, 1, 1, −1, 1, 1, −1, −1









Logical zeros become weighted by an arithmetic value, −1. In this manner the optimum correlator trajectory for a particular chip sequence is projected through the correlator trellis. The example above corresponds to the in-phase correlator waiting for an originally transmitted data sequence d0 . . . d7 of 0,0,1,0,1,0,1,0. For this example, that particular branch represents a correlation provided in Eq. 11;

y42=x0(−1)+x1(1)+x2(1)+x3(−1)+x4(1)+x5(1)+x6(−1)+x7(−1)   (Eq. 11)

x0 . . . x7 represent corrupted or noisy input signal samples. When the xi match the coefficients significantly then that 8-tuple (1 of 64) output possesses maximum weighting and is declared most, likely by the magnitude comparator. Another strategy seeks to minimize the distance between the xi and ci. The process is similar in either case.
Table 2 illustrates example in-phase and quadrature 4-tuple combinations. It is noted that the examples provided in Tables 1 and 2 are provided for illustrative purposes and are not limiting. Other specific examples will be apparent to persons skilled in the relevant arts based on the teachings herein, and such other examples are within the scope and spirit of the invention.












TABLE 2





d0 d1 d2 d3

4-tuple

4-tuple



d4 d5 d6 d7
In phase
Combination
Quadrature
Combination
Complex





















D0
00000000
11101101
B2 + B20
11101101
B2 + B20
111−111−11


D1
00000001
00011101
B13 + B20
11101101
B2 + B20
jjj−j11−11


D2
00000010
00011101
B13 + B20
00011101
B13 + B20 
−1−1−1111−11


D3
00000011
11101101
B2 + B20
00011101
B13 + B20
−j−j−jj11−11


D4
00000100
00100001
B14 + B23
11101101
B2 + B20
jj1−1jj−11


D5
00000101
00010001
B13 + B23
00101101
B14 + B20
−1−1j−jjj−11


D6
00000110
11010001
B1 + B23
00011101
B13 + B20
−j−j−11jj−11


D7
00000111
11100001
B2 + B23
11011101
B1 + B20
11−jjjj−11


D8
00001000
00100001
B14 + B23
00100001
B14 + B23
−1−11−1−1−1−11


D9
00001001
11010001
B1 + B23
00100001
B14 + B23
−j−jj−j−1−1−11


D10
00001010
11010001
B1 + B23
11010001
B1 + B23
11−11−1−1−11


D11
00001011
00100001
B14 + B23
11010001
B1 + B23
jj−jj−1−1−11


D12
00001100
11101101
B2 + B20
00100001
B14 + B23
−j−j1−1−j−j−11


D13
00001101
11011101
B1 + B20
11100001
B2 + B23
11j−j−j−j−11


D14
00001110
00011101
B13 + B20
11010001
B1 + B23
jj−11−j−j−11


D15
00001111
00101101
B14 + B20
00010001
B13 + B23
−1−1−jj−j−j−11


D16
00010000
01000111
B7 + B17
11101101
B2 + B20
j1j−1j1−j1


D17
00010001
00010111
B13 + B17
01001101
B7 + B20
−1j−1−jj1−j1


D18
00010010
10110111
B8 + B17
00011101
B13 + B20
−j−1−j1j1−j1


D19
00010011
11100111
B2 + B17
10111101
B8 + B20
1−j1jj1−j1


D20
00010100
00000011
B15 + B19
01100101
B6 + B21
−1jj−1−1j−j1


D21
00010101
10010011
B9 + B19
00000101
B15 + B21
−j−1−1−j−1j−j1


D22
00010110
11110011
B0 + B19
10010101
B9 + B21
1−j−j1−1j−j1


B23
00010111
01100011
B6 + B19
11110101
B0 + B21
j11j−1j−j1


D24
00011000
10001011
B11 + B18
00100001
B14 + B23
−j−1j−1−j−1−j1


D25
00011001
11011011
B1 + B18
10000001
B11 + B23
1−j−1−j−j−1−j1


D26
00011010
01111011
B4 + B18
11010001
B1 + B23
j1−j1−j−1−j1


D27
00011011
00101011
B14 + B18
01110001
B4 + B23
−1j1j−j−1−j1


D28
00011100
11001111
B3 + B16
10101001
B10 + B22
1−jj−11−j−j1


D29
00011101
01011111
B5 + B16
11001001
B3 + B22
j1−1−j1−j−j1


D30
00011110
00111111
B12 + B16
01011001
B5 + B22
−1j−j11−j−j1


D31
00011111
10101111
B10 + B16
00111001
B12 + B22
−j−11j1−j−j1


D32
00100000
01000111
B7 + B17
01000111
B7 + B17
−11−1−1−1111


D33
00100001
10110111
B8 + B17
01000111
B7 + B17
−jj−j−j−1111


D34
00100010
10110111
B8 + B17
10110111
B8 + B17
1−111−1111


D35
00100011
01000111
B7 + B17
10110111
B8 + B17
j−jjj−1111


D36
00100100
10001011
B11 + B18
01000111
B7 + B17
−jj−1−1−jj11


D37
00100101
10111011
B8 + B18
10000111
B11 + B17
1−1−j−j−jj11


D38
00100110
01111011
B4 + B18
10110111
B8 + B17
j−j11−jj11


D39
00100111
01001011
B7 + B18
01110111
B4 + B17
−11jj−jj11


D40
00101000
10001011
B11 + B18
10001011
B11 + B18
1−1−1−11−111


D41
00101001
01111011
B4 + B18
10001011
B11 + B18
j−j−j−j1−111


D42
00101010
01111011
B4 + B18
01111011
B4 + B18
−11111−111


D43
00101011
10001011
B11 + B18
01111011
B4 + B18
−jjjj1−111


D44
00101100
01000111
B7 + B17
10001011
B11 + B18
j−j−1−1j−j11


D45
00101101
01110111
B4 + B17
01001011
B7 + B18
−11−j−jj−j11


D46
00101110
10110111
B8 + B17
01111011
B4 + B18
−jj11j−j11


D47
00101111
10000111
B11 + B17
10111011
B8 + B18
1−1jjj−j11


D48
00110000
11101101
B2 + B20
01000111
B7 + B17
−j1−j−1−j1j1


D49
00110001
10111101
B8 + B20
11100111
B2 + B17
1j1−j−j−j1


D50
00110010
00011101
B3 + B20
10110111
B8 + B17
j−1j1−j1j1


D51
00110011
01001101
B7 + B20
00010111
B3 + B17
−1−j−1j−j1j1


D52
00110100
10101001
B10 + B22
11001111
B3 + B16
1j−j−11jj1


D53
00110101
00111001
B12 + B22
10101111
B10 + B16
j−11−j1jj1


D54
00110110
01011001
B5 + B22
00111111
B12 + B16
−1−jj11jj1


D55
00110111
11001001
B3 + B22
01011111
B5 + B16
−j1−1j1jj1


D56
00111000
00100001
B14 + B23
10001011
B11 + B18
j−1−j−1j−1j1


D57
00111001
01110001
B4 + B23
00101011
B14 + B18
−1−j1−jj−1j1


D58
00111010
11010001
B1 + B23
01111011
B4 + B18
−j1j1j−1j1


D59
00111011
10000001
B11 + B23
11011011
B1 + B18
1j−1jj−1j1


D60
00111100
01100101
B6 + B21
00000011
B15 + B19
−1−j−j−1−1−jj1


D61
00111101
11110101
B0 + B21
01100011
B6 + B19
−j11−j−1−jj1


D62
00111110
10010101
B9 + B21
11110011
B0 + B19
1jj1−1−jj1


D63
00111111
00000101
B15 + B21
10010011
B9 + B19
j−1−1j−1−jj1









IX CCK Decoder

A. Introduction
A fast correlator transform (FCT) kernel in accordance with the invention, can be used as a building block for a CCK decoder. For example, a FCT kernal can be applied and used to decode data using the IEEE 802.11b 11 Mbps signaling scheme. For the design, two correlators are preferably used. Since the signaling scheme is complex one correlator is used for the in-phase (I) channel, and another correlator is used for the quadrature phase (Q) channel. The input to each channel is the designated codeword for the transmitted data. Table 3 illustrates example input data and corresponding coded output.







TABLE 3







Input Data and CCK Coded Output











Quadrature


Data (input)
Inphase (output)
(output)





00000001
00011101 (B13 + B20)
11101101




(B2 + B20)









The CCK coded data for each channel is passed into its respective correlator for processing. The output of a branch correlator trellis preferably has a maximum value of 8 (with no noise in the system) that corresponds to a correlation with the input codeword. The value can be a ∓8 depending on the phase. The value is then magnitude squared, which will give one branch of the trellis a maximum value of 64. The in-phase and quadrature branches of the I and Q correlators are then paired together according to the CCK codeword in Table 2, above. These pairs, when combined, provide a maximum I2+Q2 value of 128.
The CCK decoder works fine when there is a 0 or π phase rotation. A problem arises, however, when there is a π/2 or a 3π/2 phase rotation. The problem is described with the following example.
The base codeword Table 2, above, for the 11 Mbps CCK encoding scheme gives the pairs of codewords that correspond to a given input message. For 56 out of the possible 64 codewords, the encoded words on the I and Q channel are similar, except that they are on opposite channels. An example of two codewords are shown in Table 4 below.







TABLE 4







CCK Input and Encoded Data









Data (input)
Inphase (output)
Quadrature (output)





00001100
11101101 (B2 + B20) 
00100001 (B14 + B23)


00000100
00100001 (B14 + B23)
11101101 (B2 + B20) 









The codewords for the two data symbols given in Table 4 are similar, but are on different channels. So if there is a π/2 or a 3π/2 phase shift from differentially encoding the transmitted data, then the I and Q channels will be swapped. This is illustrated in FIG. 9, which is a signal path diagram for an example CCK decoder output trellis, including an I signal path 902 and a Q signal path 904.
If the codewords received on the I signal path 902 and the Q signal path 904 are 11101101 and 00100001, respectively, then the maximum output will be 8 and will follow paths 906a and 906b, respectively. But if the codewords received on the I signal path 902 and the Q signal path 904 are swapped, then the maximum output will be 8 and will follow paths 908a and 908b, respectively, which correspond to different input data than paths 906a and 906b. This is acceptable if that is the intended path. But if there is a π/2 or a 3π/2 phase rotation, then the input words to the correlator are swapped on the I and Q channels, and the I or Q channel is inverted depending on what the phase rotation was. For these cases, if the input codewords before differential encoding correspond to the paths 906a and 906b, then the decoded data should be that which is associated with the paths 906a and 906b, not the paths 908a and 908b. Since the differential decoding process is typically performed after or during the correlation process, then the phase rotated symbols enter the correlator in that form. A determination should be made as to whether there was a phase rotation. Phase rotation can be discerned as follows. If there was a π/2 phase shift, the in-phase channel will be inverted. If there is a 3π/2 phase shift, the quadrature phase channel will be inverted.
FIG. 10 is a block diagram of an inverted channel detection circuit 1000 that can be used to determine if one of the channels was inverted. If both channels were inverted then there was a π phase shift and the data is decoded correctly, so this is of no consequence to the system.
X. CCK Correlator Optimizations
A. Introduction
A CCK parallel correlator in accordance with the invention can be optimized to reduce the number of adders in a Fast Correlator Transform (FCT) kernel. An initial complex correlation is presented below, followed by various optional optimizations that reduce the complexity of the initial complex correlation.
Table 6 lists 64 complex codewords that have coefficients with four possible values of +1, −1, +j, and −j. This means that either the real part or imaginary part of each coefficient is 0, but not both. Table 6also lists corresponding codewords rotated by 45° to get the I,Q representation. Note that each coefficient in the I,Q representation has a magnitude of √{square root over (2)} since the four possible coefficients are now +1+j, +1−j, −1+j, and −1−j. Rotation of the coefficients is the key to simplification of the FCT.
Additional simplification is achieved by realizing that some intermediate results are the negative of other intermediate results at the same stage.
B. Initial Complex Correlation
An initial complex correlation of complex CCK codewords with complex inputs can be achieved from equation 12:




  
    
      
        
          
            
              
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                            )
                          
                        
                      
                    
                    -
                    
                      
                        ∑
                        
                          k
                          =
                          0
                        
                        7
                      
                      ⁢
                      
                        
                          
                            c
                            Q
                          
                          ⁡
                          
                            (
                            k
                            )
                          
                        
                        ⁢
                        
                          
                            x
                            I
                          
                          ⁡
                          
                            (
                            
                              n
                              -
                              k
                            
                            )
                          
                        
                      
                    
                  
                  ]
                
              
            
          
        
      
    
    
      
        (
        
          Eq
          .
          
            
          
          ⁢
          13
        
        )
      
    
  




FIG. 11 is a block diagram 1100 of the complex correlation of equation 12.
The resulting complex correlation of equation 12 includes four real correlates performed in parallel. Implementing the correlate equations of equation 12 results in 4×7+2=30 additions per codeword. For the 64 CCK codewords there are 64×30=1920 additions to implement all the required correlations. Since the real or imaginary part of each CCK complex coefficient is zero, the computation is reduced to 1920−(64×2×7)=1024.
C. Fast Correlator Transform (FCT)
A Fast Correlator Transform (FCT) can be used to reduce the computation by assuming the use of 45° rotated I,Q codewords shown in Table 6, below. To derive the 64 parallel correlates for all of the codewords, two substantially similar FCTs are preferred. One is for the real part of the input. The other is for the imaginary part of the input. FIG. 12 is a block diagram of an example FCT 1200. For a CCK with codeword size of 64 the FCT 1200 can be divided into several stages. For example, FIG. 13 illustrates a FCT 1300 that includes stages 1a (1302), 1b (1304), 2a (1306), 2b (1308), and 3 (1310), which are described below.
FIG. 14 is an example block diagram of stage 1a of the FCT 1300. In the example of FIG. 14, the stage 1a includes 2 additions and 6 subtractions, which implement the following:

D0=x0+x1, D1=−x0+x1, D2=x0−x1, D3=−x0−x1 

D4=x2+x3, D5=−x2+x3, D6=x2−x3, D7=−x2−x3.

FIG. 15 is an example block diagram of stage 1b of the FCT 1300. In the example of FIG. 15, the stage 1b includes 8 additions and subtractions, which implement the following:

D8=x4+x5, D9=−x4+x5, D10=x4−x5, D11=−x4−x5 

D12=x6+x7, D13=−x6+x7, D14=x6−x7, D15=−x6−x7.

FIG. 16 is an example block diagram of stage 2a of the FCT 1300. In the example of FIG. 16, the stage 2a includes 16 additions, which implement the following:

B0=D0+D4, B1=D0+D5, B2=D0+D6, B3=D0+D7 

B4=D1+D4, B5=D1+D5, B6=D1+D6, B7=D1+D7 

B8=D2+D4, B9=D2+D5, B10=D2+D6, B11=D2+D7 

B12=D3+D4, B13=D3+D5, B14=D3+D6, B15=D3+D7.

FIG. 17 is an example block diagram of stage 2b of the FCT 1300. In the example of FIG. 17, the stage 2b includes 16 additions, which implement the following:

B16=D8+D12, B17=D9+D12, B18=D10+D12, B19=D11+D12 

B20=D8+D13, B21=D9+D13, B22=D10+D13, B23=D11+D13 

B24=D8+D14, B25=D9+D14, B26=D10+D14, B27=D11+D14 

B28=D8+D15, B29=D9+D15, B30=D10+D15, B31=D11+D15 

Stage 3 of the FCT 1300 includes 64 additions for the I values and 64 additions for the Q values, as follows:

I0=B2+B20, I1=B13+B20, I2=B13+B20, I3=B2+B20 

Q0=B2+B20, Q1=B2+B20, Q2=B13+B20, Q3=B13+B20 

I4=B14+B23, I5=B13+B23, I6=B1+B23, I7=B2+B23 

Q4=B2+B20, Q5=B14+B20, Q6=B13+B20, Q7=B1+B20 

I8=B14+B23, I9=B1+B23, I10=B1+B23, I11=B14+B23 

Q8=B14+B23, Q9=B14+B23, Q10=B1+B23, Q11=B1+B23 

I12=B2+B20, I13=B1+B20, I14=B13+B20, I15=B14+B20 

Q12=B14+B23, Q13=B2+B23, Q14=B1+B23, Q15=B13+B23 

I16=B7+B17, I17=B13+B17, I18=B8+B17, I19=B2+B17 

Q16=B2+B20, Q17=B7+B20, Q18=B13+B20, Q19=B8+B20 

I20=B15+B19, I21=B9+B19, I22=B0+B19, I23=B6+B19 

Q20=B6+B21, Q21=B15+B21, Q22=B9+B21, Q23=B0+B21 

I24=B11+B18, I25=B1+B18, I26=B4+B18, IB14=B18 

Q24=B14+B23, Q25=B11+B23, Q26=B1+B23, Q27=B4+B23 

I28=B3+B16, I29=B5B16, I30=B12+B16, I31=B10+B16 

Q28=B10+B22, Q29=B3+B22, Q30=B5+B22, Q31=B12+B22 

I32=B7+B17, I33=B8+B17, I34=B8+B17, I35=B7+B17 

Q32=B7+B17, Q33=B7+B17, Q34=B8+B17, Q35=B8+B17 

I36=B11+B18, I27=B8+B18, I38=B4+B18, I39=B7+B18 

Q36=B7+B17, Q37=B11+B17, Q38=B8+B17, Q39=B4+B17 

I40=B11+B18, I41=B4+B18, I42=B4+B18, I43=B11+B18 

Q40=B11+B18, Q41=B11+B18, Q42=B4+B18, Q43=B4+B18 

I44=B7+B17, I45=B4+B17, I46=B11+B17, I47=B11+B17 

Q44=B11+B18, Q45=B7+B18, Q46=B4+B18, Q47=B8+B18 

I48=B2+B20, I49=B8+B20, I50=B13+B20, I51=B7+B20 

Q48=B7+B17, Q49=B2+B17, Q50=B8+B17, Q51=B13+B17 

I52=B10+B22, I53=B12+B22, I54=B5+B22, I55=B3+B22 

Q52=B3+B16, Q53=B10+B16, Q54=B12+B16, Q55=B5+B16 

I56=B14+B23, I57=B4+B23, I58=B1+B22, I59=B11+B23 

Q56=B11+B18, Q57=B14+B18, Q58=B4+B18, Q59=B1+B18 

I60=B6+B21, I61=B0+B21, I62=B9+B21, I62=B15+B21 

Q60=B15+B19, Q61=B6+B19, Q62=B0+B19, Q63=B9+B19 

Stage 3 can be optimized by noting that there are 40 distinct values for I and Q which are

A0=B2+B20, A1=B13+B20, A2=B14+B23, A3=B13+B23 

A4=B14+B20, A5=B1+B23, A6=B2+B23, A7=B1+B20 

A8=B7+B17, A9=B13+B17, A10=B7+B20, A11=B8+B17 

A12=B2+B17, A13=B8+B20, A14=B15+B19, A15=B6+B21 

A16=B9+B19, A17=B15+B21, A18=B0+B19, A19=B9+B21 

A20=B6+B19, A21=B0+B21, A22=B11+B18, A23=B1+B18 

A24=B11+B23, A25=B4+B18, A26=B14+B18, A27=4+B23 

A28=B3+B16, A29=B10+B22 A30=B5+B16, A31=B3+B22 

A32=B12+B16, A33=B5+B22, A34=B10+B16, A35=B12+B22 

A36=B8+B18, A37=B11+B17, A38=B7+B18, A39=B4+A17.

FIG. 18 is a block diagram of an example system 1800 that calculates the 40 An values. The 40 An values are used to calculate the final 64 output values and are described below.
Further optimization of the FCT 1300 can be achieved by considering the relationship:

Bn=−B15-n,8≦n≦15   (Eq. 14)

The relationship of equation 14 eliminates the need to calculate the eight values of Bn for 8≦n≦15, and the intermediate results D2 and D3. Thus, 8+2=10 additions can be removed and be replaced by 10 negates of the samples, or subtraction instead of addition at the next processing stage.
Because c7 is always 1, calculation of B24 . . . B31 can eliminate the following eight additions:

B24=D8+D14, B25=D9+D14, B26=D10+D14, B27=D11+D14 

B28=D8+D15, B29=D9+D15, B30=D10+D15, B31=D11+D15 

and, consequently, the following two additions:

D14=x6−x7, D15=−x6−x7.

Four more additions can be removed by considering that:

D6=−D5, D7=−D4, D10=−D9, D11=−D8 

Table 6 shows the 64 correlate outputs as a function of B0 . . . B7 and B16 . . . B23. Note that 40 distinct summations, or subtractions, are used to calculate the 128 results in the final stage. FIG. 19 is a block diagram of connection stages 1a, 1b, 2a, 2b, and 3 for an example simplified architecture FCT 1900. Note the reduction in states between stages from the FCT 1300 in FIG. 13 and the FCT 1900 in FIG. 19. The FCT 1900 includes stages 1a (1902), 1b (1904), 2a (1906), 2c (1908), and 3 (1910), which are described below.
FIG. 20 is an example block diagram of the stage 1a of the FCT 1900. In the example of FIG. 20, stage 1a includes 4 additions and subtractions, which implement the following:

D0=x0+x1, D1=31 x0+x1 

D4=x2+x3, D5=−x2+x3 

FIG. 21 is an example block diagram of the stage 1b of the FCT 1900. In the example of FIG. 21, stage 1b includes 4 additions or subtractions, which implement the following:

D8=x4+x5, D9=−x4+x5 

D12=x6+x7, D13=−x6+x7 

FIG. 22 is an example block diagram of stage 2a of the FCT 1900. In the example of FIG. 22, stage 2a includes 8 additions or subtractions, which implement the following:

B0=D0+D4, B1=D0+D5, B2=D0−D5, B3=D0−D4 

B4=D1+D4, B5=D1+D5, B6=D1−D5, B7=D1−D4 

FIG. 23 is an example block diagram of stage 2b of the FCT 1900. In the example of FIG. 23, stage 2b includes 8 additions, which implement the following:

B16=D8+D12, B17=D9+D12, B18=−D9+D12, B19=−D8+D12 

B20=D8+D13, B21=D9+D13, B22=−D9+D13, B23=−D8+D13 

FIGS. 24A and 24B illustrate an example block diagram of stage 3 of the FCT 1900. In the example of FIGS. 24A and 24B, stage 3 includes 40 additions or subtractions, which implement the following:

A0=B2+B20, A1=−B2+B20, A2=B1+B23, A3=B2+B23 

A4=−B1+B20, A5=B1+B23, A6=B2+B23, A7=B1+B20 

A8=B7+B17, A9=−B2+B17, A10=B7+B20, A11=−B7+B17 

A12=B2+B17, A13=−B7+B20, A14=−B0+B19, A15=B6+B21 

A16=−B6+B19, A17=−B0+B21, A18=B0+B19, A19=−B6+B21 

A20=B6+B19, A21=B0+B21, A22=−B4+B18, A23=B1+B18 

A24=−B4+B23, A25=B4+B18, A26=−B1+B18, A27=B4+B23 

A28=B3+B16, A29=−B5+B22 A30=B5+B5+B16, A31=B3+B22 

A32=−B3+B16, A33=B5+B22, A34=−B5+B16, A35=−B3+B22 

A36=−B7+B18, A37=−B4+B17, A38=B7+B18, A39=B4+B17 

Final outputs are calculated using 128 additions or subtractions.

YI0=AI0+AQ0, YQ0=AQ0−AI0, YI1=AI1+AQ0, YQ1=AQ1−AI0 

YI2=AI1+AQ1, YQ2=AQ1−AI1, YI3=AI0+AQ1, YQ3=AQ0−AI1 

YI4=AI2+AQ0, YQ4=AQ2−AI0, YI5=AI3+AQ4, YQ5=AQ3−AI4 

YI6=AI5+AQ1, YQ6=AQ5−AI1, YI7=AI6+AQ7, YQ7=AQ6−AI7 

YI8=AI2+AQ2, YQ8=AQ2−AI2, YI9=AI5+AQ2, YQ9=AQ5−AI2 

YI10=AI5+AQ5, YQ10=AQ5−AI5, YI11=AI2+AQ5, YQ11=AQ2−AI5 

YI12=AI0+AQ2, YQ12=AQ0−AI2, YI13=AI7+AQ6, YQ13=AQ7−AI6 

YI14=AI1+AQ5, YQ14=AQ1−AI5, YI15=AI4+AQ3, YQ15=AQ4−AI3 

YI16=AI8+AQ0, YQ16=AQ8−AI0, YI17=AI9+AQ10, YQ17=AQ9−AI10 

YI18=AI11+AQ1, YQ18=AQ11−AI1, YI19=AI12+AQ13, YQ19=AQ12−AI13 

YI20=AI14+AQ15, YQ20=AQ14−AI15, YI21=AI16+AQ17, YQ21=AQ16−AI17 

YI22=AI18+AQ19, YQ22=AQ18−AI19, YI23=AI20+AQ21, YQ23=AQ20−AI21 

YI24=AI22+AQ2, YQ24=AQ22−AI2, YI25=AI23+AQ24, YQ25=AQ23−AI24 

YI26=AI25+AQ5, YQ26=AQ25−AI5, YI27=AI26+AQ27, YQ27=AQ26−AI27 

YI28=AI28+AQ29, YQ28=AQ28−AI29, YI29=AI30+AQ31, YQ29=AQ30−AI31 

YI30=AI32+AQ33, YQ30=AQ32−AI33, YI31=AI34+AQ35, YQ31−AQ34−AI35 

YI32=AI8+AQ8, YQ32=AQ8−AI8, YI33=AI11+AQ8, YQ33=AQ11−AI8 

YI34=AI11+AQ11, YQ34=AQ11−AI11, YI35=AI8+AQ11, YQ35=AQ8−AI11 

YI36=AI22+AQ8, YQ36=AQ22−AI8, YI37=AI36+AQ37, YQ37=AQ36−AI37 

YI38=AI25+AQ11, YQ38=AQ25−AI11, YI39=AI38+AQ39, YQ39=AQ38−AI39 

YI40=AI22+AQ22, YQ40=AQ22−AI22, YI41=AI25+AQ22, YQ41=AQ25−AI22 

YI42=AI25+AQ25, YQ42=AQ25−AI25, YI43=AI22+AQ25, YQ43=AQ22−AI25 

YI44=AI8+AQ22, YQ44=AQ8−AI22, YI45=AI39+AQ38, YQ45=AQ39−AI38 

YI46=AI1+AQ25, YQ46=AQ11−AI25, YI47=AI37+AQ36, YQ47=AQ37−AI36 

YI48=AI0+AQ8, YQ48=AQ0−AI8, YI49=AI13+AQ12, YQ49=AQ13−AI12 

YI50=AI1+AQ11, YQ50=AQ1−AI11, YI51=AI10+AQ9, YQ51=AQ10−AI9 

YI52=AI29+AQ28, YQ52−AQ29−A28, YI53=AI35+AQ34, YQ53=AQ35−A34 

YI54=AI33+AQ32, YQ54=AQ33−A32, YI55=AI31+AQ30, YQ55=AQ31−A30 

YI56=AI2+AQ22, YQ56=AQ2−A22, YI57=AI27+AQ26, YQ57=AQ27−A26 

YI58=AI5+AQ25, YQ58=AQ5−A25, YI59=AI24+AQ23, YQ59=AQ24−A23 

YI60=AI15+AQ14, YQ60=AQ15−AI4, YI61=AI21+AQ20 , YQ61=AQ21−A20 

YI62=AI19+AQ18, YQ62=AQ19−AI8, YI63=AI17+AQ16, YQ63=AQ17−AI6 

The number of adders in the final stage can be reduced by almost half from 128 to, 72 by sharing adders. The adders are shared between codewords that are complex conjugates pairs as listed in Table 5. There are a total of 36 complex conjugate pairs where 8 of the codewords are complex conjugates of themselves (real coefficients). Consider, as an example, the pair of codeword 1 and codeword 3, from Table 5, with the following output equations:

YI1=AI1+AQ0, YQ1=AQ1−AI0   (Eq. 15)

YI3=AI0+AQ1, YQ3=AQ0−AI1   (Eq. 16)

The four additions, or subtractions, can be reduced to two additions by using the architecture shown in FIG. 18. A control signal sets the branches shown to opposite signs and the registers are loaded on opposite edges of the clock. Outputs include four output values stored in the four registers. This structure is repeated 36 times for each of the conjugate a, b codeword pairs in Table 5.
To summarize, the total number of complex additions, or subtractions, for the simplified parallel correlator 1900 are:




  
    
      
        
          Stage
          ⁢
          
            
          
          ⁢
          #1
        
        =
        8
      
    
  
  
    
      
        
          Stage
          ⁢
          
            
          
          ⁢
          #2
        
        =
        16
      
    
  
  
    
      
        
          
            Stage
            ⁢
            
              
            
            ⁢
            #3
          
          =
          
            
              40
              +
              36
            
            =
            76
          
        
        _
      
    
  
  
    
      
        TOTAL
        =
        100
      
    
  











TABLE 5







Codeword Conjugate Pairs










Codeword
Codeword



Index a
Index b














0
—



1
3



2
—



4
12



5
15



6
14



7
13



8
—



9
11



10
—



16
48



17
51



18
50



19
49



20
60



21
63



22
62



23
61



24
56



25
59



26
58



27
57



28
52



29
55



30
54



31
53



32
—



33
35



34
—



36
44



37
47



38
46



39
45



40
—



41
43



42
—










D. FCT Based on CCK Code Properties
Another less complex embodiment can be derived by considering the equations for generating the CCK correlation output:

d=c7x7+c6x6+c5x5+c4x4+c3x3+c2x2+c1x1+c0x0   (Eq. 16)

where c7,c6,c5,c4,c3,c2,c1,c0 are the complex coefficients and x7,x6,x5,x4,x3,x2,x1,x0 are the complex input buffer samples. The codewords can be uniquely defined by the three coefficients, c6,c5,c3, with the other coefficients defined by

c0=c6c5c3 

c1=c5c3 

c2=c6c3 

c4=−c6c5 

c7=1

Thus, after substituting coefficient relationships and rearranging, the correlation output becomes:

d=x7+c6x6+c5(x5−c6x4)+c3{x3+c6x2+c5[x1+c6x0]}  (Eq. 17)

For correlation with a single codeword, the resulting structure for equation 170 is shown in FIG. 25.
The FCT is derived from the single correlate by converting each coefficient branch, starting at the input, to a positive and negative branch. The resulting structure is shown in FIGS. 26 and 27.
Stage 1 consists of 4 additions or subtractions.

D0=x0+x1, D1=−x0+x1, D2=x2, D3=x3 

D4=x4+x5, D5=−x4+x5, D6=x6, D7=x7 

Stage 2 consists of 8 additions or subtractions.

E0=D0+D2, E1=D2−D0, E2=D1+D2, E3=D1−D2 

E4=D4+D6, E5=D6−D4, E6=D5+D6, E7=D5−D6 

Stage 3 consists of 16 additions or subtractions.

B0=E0+D3, −B3=E1+D3, −B2=−E0+D3, B1=−E1+D3 

B4=E2+D3, B5=E3+D3, −B6=−E2+D3, −B7=−E3+D3 

B16=E4+D7, B18=E5+D7, B23=−E4+D7, B20=−E5+D7 

B17=E6+D7, B21=E7+D7, B22=−E6+D7, B18=−E7+D7 

Stage 4 consists of 40 additions or subtractions.

A0=B2+B20, A1=−B2+B20, A2=−B1+B23, A3=−B2+B23 

A4=−B1+B20, A5=B1+B23, A6=B2+B23, A7=B1+B20 

A8=B7+B17, A9=−B2+B17, A10=B7+B20, A11=B7+B17 

A12=B2+B17, A13=−B7+B20, A14=−B0+B19, A15=B6+B21 

A16=−B6+B19, A17=−B0+B21, A18=B0+B19, A19=−B6+B21 

A20=B6+B19, A21=B0+B21, A22=−B4+B18, A23=B1+B18 

A24=−B4+B23, A25=B4+B18, A26=−B1+B18, A27=B4+B23 

A28=B3+B16, A29=−B5+B22 A30=B5+B16, A31=B3+B22 

A32=−B3+B16, A33=B5+B22, A34=−B5+B16, A35=−B3+B22 

A36=−B7+B18, A37=−B4+B17, A38=B7+B18, A39=B4+B17 












TABLE 6





Index
Bits
Complex Codeword
I, Q Codeword
I, Q Combo



















0
000000
+1+1+1−1+1+1−1+1
+1+1+1−1+1+1−1+1
B2 + B20





+1+1+1−1+1+1−1+1
B2 + B20


1
000001
+j+j+j−j+1+1−1+1
−1−1−1+1+1+1−1+1
−B2 + B20





+1+1+1−1+1+1−1+1
B2 + B20


2
000010
−1−1−1+1+1+1−1+1
−1−1−1+1+1+1−1+1
−B2 + B20





−1−1−1+1+1+1−1+1
−B2 + B20


3
000011
−j−j−j+j+1+1−1+1
+1+1+1−1+1+1−1+1
B2 + B20





−1−1−1+1+1+1−1+1
−B2 + B20


4
000100
+j+j+1−1+j+j−1+1
−1−1+1−1−1−1−1+1
−B1 + B23





+1+1+1−1+1+1−1+1
B2 + B20


5
000101
−1−1+j−j+j+j−1+1
−1−1−1+1−1−1−1+1
−B2 + B23





−1−1+1−1+1+1−1+1
−B1 + B20


6
000110
−j−j−1+1+j+j−1+1
+1+1−1+1−1−1−1+1
B1 + B23





−1−1−1+1+1+1−1+1
−B2 + B20


7
000111
+1+1−j+j+j+j−1+1
+1+1+1−1−1−1−1+1
B2 + B23





+1+1−1+1+1+1−1+1
B1 + B20


8
001000
−1−1+1−1−1−1−1+1
−1−1+1−1−1−1−1+1
−B1 + B23





−1−1+1−1−1−1−1+1
−B1 + B23


9
001001
−j−j+j−j−1−1−1+1
+1+1−1+1−1−1−1+1
B1 + B23





−1−1+1−1−1−1−1+1
−B1 + B23


10
001010
+1+1−1+1−1−1−1+1
+1+1−1+1−1−1−1+1
B1 + B23





+1+1−1+1−1−1−1+1
B1 + B23


11
001011
+j+j−j+j−1−1−1+1
−1−1+1−1−1−1−1+1
−B1 + B23





+1+1−1+1−1−1−1+1
B1 + B23


12
001100
−j−j+1−1−j−j−1+1
+1+1+1−1+1+1−1+1
B2 + B20





−1−1+1−1−1−1−1+1
−B1 + B23


13
001101
+1+1+j−j−j−j−1+1
+1+1−1+1+1+1−1+1
B1 + B20





+1+1+1−1−1−1−1+1
B2 + B23


14
001110
+j+j−1+1−j−j−1+1
−1−1−1+1+1+1−1+1
−B2 + B20





+1+1−1+1−1−1−1+1
B1 + B23


15
001111
−1−1−j+j−j−j−1+1
−1−1+1−1+1+1−1+1
−B1 + B20





−1−1−1+1−1−1−1+1
−B2 + B23


16
010000
+j+1+j−1+j+1−j+1
−1+1−1−1−1+1+1+1
B7 + B17





+1+1+1−1+1+1−1+1
B2 + B20


17
010001
−1+j−1−j+j+1−j+1
−1−1−1+1−1+1+1+1
−B2 + B17





−1+1−1−1+1+1−1+1
B7 + B20


18
010010
−j−1−j+1+j+1−j+1
+1−1+1+1−1+1+1+1
−B7 + B17





−1−1−1+1+1+1−1+1
−B2 + B20


19
010011
+1−j+1+j+j+1−j+1
+1+1+1−1−1+1+1+1
B2 + B17





+1−1+1+1+1+1−1+1
−B7 + B20


20
010100
−1+j+j−1−1+j−j+1
−1−1−1−1−1−1+1+1
−B0 + B19





−1+1+1−1−1+1−1+1
B6 + B21


21
010101
−j−1−1−j−1+j−j+1
+1−1−1+1−1−1+1+1
−B6 + B19





−1−1−1−1−1+1−1+1
−B0 + B21


22
010110
+1−j−j+1−1+j−j+1
+1+1+1+1−1−1+1+1
B0 + B19





+1−1−1+1−1+1−1+1
−B6 + B21


23
010111
+j+1+1+j−1+j−j+1
−1+1+1−1−1−1+1+1
B6 + B19





+1+1+1+1−1+1−1+1
B0 + B21


24
011000
−j−1+j−1−j−1−j+1
+1−1−1−1+1−1+1+1
−B4 + B18





−1−1+1−1−1−1−1+1
−B1 + B23


25
011001
+1−j−1−j−j−1−j+1
+1+1−1+1+1−1+1+1
B1 + B18





+1−1−1−1−1−1−1+1
−B4 + B23


26
011010
+j+1−j+1−j−1−j+1
−1+1+1+1+1−1+1+1
B4 + B18





+1+1−1+1−1−1−1+1
B1 + B23


27
011011
−1+j+1+j−j−j−j+1
−1−1+1−1+1−1+1+1
−B1 + B18





−1+1+1+1−1−1−1+1
B4 + B23


28
011100
+1−j+j−1+1−j−j+1
+1+1−1−1+1+1+1+1
B3 + B16





+1−1+1−1+1−1−1+1
−B5 + B22


29
011101
+j+1−1−j+1−j−j+1
−1+1−1+1+1+1+1+1
B5 + B16





+1+1−1−1+1−1−1+1
B3 + B22


30
011110
−1+j−j+1+1−j−j+1
−1−1+1+1+1+1+1+1
−B3 + B16





−1+1−1+1+1−1−1+1
B5 + B22


31
011111
−j−1+1+j+1−j−j+1
+1−1+1−1+1+1+1+1
−B5 + B16





−1−1+1+1+1−1−1+1
−B3 + B22


32
100000
−1+1−1−1−1+1+1+1
−1+1−1−1−1+1+1+1
B7 + B17





−1+1−1−1−1+1+1+1
B7 + B17


33
100001
−j+j−j−j−1+1+1+1
+1−1+1+1−1+1+1+1
−B7 + B17





−1+1−1−1−1+1+1+1
B7 + B17


34
100010
+1−1+1+1−1+1+1+1
+1−1+1+1−1+1+1+1
−B7 + B17





+1−1+1+1−1+1+1+1
−B7 + B17


35
100011
+j−j+j+j−1+1+1+1
−1+1−1−1−1+1+1+1
B7 + B17





+1−1+1+1−1+1+1+1
−B7 + B17


36
100100
−j+j−1−1−j+j+1+1
+1−1−1−1+1−1+1+1
−B4 + B18





−1+1−1−1−1+1+1+1
B7 + B17


37
100101
+1−1−j−j−j+j+1+1
+1−1+1+1+1−1+1+1
−B7 + B18





+1−1−1−1−1+1+1+1
−B4 + B17


38
100110
+j−j+1+1−j+j+1+1
−1+1+1+1+1−1+1+1
B4 + B18





+1−1+1+1−1+1+1+1
−B7 + B17


39
100111
−1+1+j+j−j+j+1+1
−1+1−1−1+1−1+1+1
B7 + B18





−1+1+1+1−1+1+1+1
B4 + B17


40
101000
+1−1−1−1+1−1+1+1
+1−1−1−1+1−1+1+1
−B4 + B18





+1−1−1−1+1−1+1+1
−B4 + B18


41
101001
+j−j−j−j+1−1+1+1
−1+1+1+1+1−1+1+1
B4 + B18





+1−1−1−1+1−1+1+1
−B4 + B18


42
101010
−1+1+1+1+1−1+1+1
−1+1+1+1+1−1+1+1
B4 + B18





−1+1+1+1+1−1+1+1
B4 + B18


43
101011
−j+j+j+j+1−1+1+1
+1−1−1−1+1−1+1+1
−B4 + B18





−1+1+1+1+1−1+1+1
B4 + B18


44
101100
+j−j−1−1+j−j+1+1
−1+1−1−1−1+1+1+1
B7 + B17





+1−1−1−1+1−1+1+1
−B4 + B18


45
101101
−1+1−j−j+j−j+1+1
−1+1+1+1−1+1+1+1
B4 + B17





−1+1−1−1+1−1+1+1
B7 + B18


46
101110
−j+j+1+1+j−j+1+1
+1−1+1+1−1+1+1+1
−B7 + B17





−1+1+1+1+1−1+1+1
B4 + B18


47
101111
+1−1+j+j+j−j+1+1
+1−1−1−1−1+1+1+1
−B4 + B17





+1−1+1+1+1−1+1+1
−B7 + B18


48
110000
−j+1−j−1−j+1+j+1
+1+1+1−1+1+1−1+1
B2 + B20





−1+1−1−1−1+1+1+1
B7 + B17


49
110001
+1+j+1−j−j+1+j+1
+1−1+1+1+1+1−1+1
−B7 + B20





+1+1+1−1−1+1+1+1
B2 + B17


50
110010
+j−1+j+1−j+1+j+1
−1−1−1+1+1+1−1+1
−B2 + B20





+1−1+1+1−1+1+1+1
−B7 + B17


51
110011
−1−j−1+j−j+1+j+1
−1+1−1−1+1+1−1+1
B7 + B20





−1−1−1+1−1+1+1+1
−B2 + B17


52
110100
+1+j−j−1+1+j+j+1
+1−1+1−1+1−1−1+1
−B5 + B22





+1+1−1−1+1+1+1+1
B3 + B16


53
110101
+j−1+1−j+1+j+j+1
−1−1+1+1+1−1−1+1
−B3 + B22





+1−1+1−1+1+1+1+1
−B5 + B16


54
110110
−1−j+j+1+1+j+j+1
−1+1−1+1+1−1−1+1
B5 + B22





−1−1+1+1+1+1+1+1
−B3 + B16


55
110111
−j+1−1+j+1+j+j+1
+1+1−1−1+1−1−1+1
B3 + B22





−1+1−1+1+1+1+1+1
B5 + B16


56
111000
+j−1−j−1+j−1+j+1
−1−1+1−1−1−1−1+1
−B1 + B23





+1−1−1−1+1−1+1+1
−B4 + B18


57
111001
−1−j+1−j+j−1+j+1
−1+1+1+1−1−1−1+1
B4 + B23





−1−1+1−1+1−1+1+1
−B1 + B18


58
111010
−j+1+j+1+j−1+j+1
+1+1−1+1−1−1−1+1
B1 + B23





−1+1+1+1+1−1+1+1
B4 + B18


59
111011
+1+j−1+j+j−1+j+1
+1−1−1−1−1−1−1+1
−B4 + B23





+1+1−1+1+1−1+1+1
B1 + B18


60
111100
−1−j−j−1−1−j+j+1
−1+1+1−1−1+1−1+1
B6 + B21





−1−1−1−1−1−1+1+1
−B0 + B19


61
111101
−j+1+1−j−1−j+j+1
+1+1+1+1−1+1−1+1
B0 + B21





−1+1+1−1−1−1+1+1
B6 + B19


62
111110
+1+j+j+1−1−j+j+1
+1−1−1+1−1+1−1+1
−B6 + B21





+1+1+1+1−1−1+1+1
B0 + B19


63
111111
+j−1−1+j−1−j+j+1
−1−1−1−1−1+1−1+1
−B0 + B21





+1−1−1+1−1−1+1+1
−B6 + B19









XI. Conclusion

While various embodiments of the present invention have been described above, it should be understood that they have been presented by way of example only, and not limitation. Other embodiments are possible and are covered by the invention.

## Claims

1. A method for correlating an encoded data word (X0-XM−1) with encoding coefficients (C0-CM−1), wherein each of (X0-XM−1) is represented by one or more bits and each said coefficient is represented by one or more bits, wherein each said coefficient has k possible states, wherein M is greater than 1, comprising:
(1) receiving the encoded data word (X0-XM−1);
(2) multiplying X0 with states C0(0) through C0(K−1) of said coefficient C0, thereby generating results X0C0(0) through X0C0(K−1);
(3) repeating step (1) for data bits (X1-XM−1) and corresponding said coefficients (C1-CM−1), respectively;
(4) grouping said results of steps (2) and (3) into N groups and summing combinations within each of said N groups, thereby generating a first layer of correlation results;
(5) grouping the results of step (4) and summing combinations of results within each group to generate one or more additional layers of results, and repeating this process until a final layer of results includes a correlation output for states of the set of coefficients (C0-CM−1);
(6) comparing magnitudes output of said correlation outputs, thereby identifying a most likely code encoded on said data word; and
(7) outputting the most likely code.

2. The method according to claim 1, wherein (4) and (5) comprise omitting summations that would result in invalid combinations of the encoding coefficients (C0-CM−1).

3. The method according to claim 1, further comprising performing steps (1) through (7) using substantially the same hardware for in-phase and quadrature phase components of the data word (X0-XM−1).

4. The method according to claim 1, wherein said coefficients (C0-CM−1) represent real numbers.

5. The method according to claim 1, wherein said coefficients (C0-CM−1) represent complex numbers.

6. The method according to claim 1, wherein each said coefficient (C0-CM−1) is represented by a single bit.

7. The method according to claim 1, wherein each said coefficient (C0-CM−1) is represented by multiple bits.

8. The method according to claim 1, wherein said code patterns (C0-CM−1) represent a cyclic code keying (“CCK”) code set substantially in accordance with IEEE 802.11 WLAN standard.

9. The method according to claim 8, wherein:
M equals 8;
each said coefficient (C0-CM−1) has two states, plus and minus;
N equals 4;
said first level of results comprises at least a portion of the following;


(X0C0+X1C1), (X0(—C0)+X1C1), (X0C0+X1(—C1)), (X0(—C0)+X1(—C1)), (X2C2+X3C3), (X2(—C2)+X3C3), (X2C2+X3(—C3)), (X2(—C2)+(X3(—C3)), (X4C4+X5C5), (X4(—C4)+X5C5), (X4C4+X5(—C5)), (X4(—C4)+X5(—C5)), (X6C6+X7C7), (X6(−C6)+X7C7), (X6C6+X7(—C7)), and (X6(—C6)+(X7(—C7)); and


wherein said second level of results comprises at least a portion of the following:


((X0C0+X1C1)+(X2C2+X3C3)), (i.e., B0),



((X0C0+X1C1)+(X2(—C2)+X3C3)), (i.e., B1),



((X0C0+X1C1)+(X2C2+X3(—C3)), (i.e., B2),



((X0C0+X1C1)+(X2(—C2)+(X3(—C3)), (i.e., B3),



((X0(—C0)+X1C1)+(X2C2+X3C3)), (i.e., B4)



((X0(—C0)+X1C1)+(X2(—C2)+X3C3)), (i.e., B5),



((X0(—C0)+X1C1)+(X2C2+X3(—C3)), (i.e. B6),



((X0(—C0)+X1C1)+(X2(—C2)+(X3(—C3)), (i.e., B7),



((X0C0+X1(—C1))+(X2C2+X3C3)), (i.e., B8)



((X0C0+X1(—C1))+(X2(—C2)+X3C3)), (i.e., B9),



((X0C0+X1(—C1))+(X2C2+X3(—C3)), (i.e., B10),



((X0C0+X1(—C1))+(X2(—C2)+(X3(—C3)), (i.e., B11),



((X0(—C0)+X1(—C1))+(X2C2+X3C3)), (i.e., B12)



((X0(—C0)+X1(—C1))+(X2(—C2)+X3C3)), (i.e., B13),



((X0(—C0)+X1(—C1))+(X2C2+X3(—C3)), (i.e., B14),



((X0(—C0)+X1(—C1))+(X2(—C2)+(X3(—C3)), (i.e., B15),



((X4C4+X5C5)+(X6C6+X7C7)), (i.e., B16),



((X4C4+X5C5)+(X6(—C6)+X7C7)), (i.e., B20),



((X4C4+X5C5)+(X6C6+X7(—C7)), (i.e., B24),



((X4C4+X5C5)+(X6(—C6)+(X7(—C7)), (i.e., B28),



((X4(—C4)+X5C5)+(X6C6+X7C7)), (i.e., B17)



((X4(—C4)+X5C5)+(X6(—C6)+X7C7)), (i.e., B21),



((X4(—C4)+X5C5)+(X6C6+X7(—C7)), (i.e., B25),



((X4(—C4)+X5C5)+(X6(—C6)+(X7(—C7)), (i.e., B29),



((X4C4+X5(—C5))+(X6C6+X7C7)), (i.e., B18)



((X4C4+X5(—C5))+(X6(—C6)+X7C7)), (i.e., B22),



((X4C4+X5(—C5))+(X6C6+X7(—C7)), (i.e., B26),



((X4C4+X5(—C5))+(X6(—C6)+(X7(—C7)), (i.e., B30),



((X4(—C4)+X5(—C5))+(X6C6+X7C7)), (i.e., 19)



((X4(—C4)+X5(—C5))+(X6(—C6)+X7C7)), (i.e., B23),



((X4(—C4)+X5(—C5))+(X6C6+X7(—C7)), (i.e., B27),



((X4(—C4)+X5(—C5))+(X6(—C6)+(X7(—C7)), (i.e., B31).

10. The method according to claim 9, wherein said second level of results omits one or more of B0 through B31.

11. The method according to claim 9, wherein said second level of results omits one or more of B0 through B31 that represent invalid combinations of one or more of (C0-CM−1).

12. The method according to claim 9, wherein said second level of results omits one or more of B0 through B31 where the omitted combination(s) would be redundant based on said CCK code specification.

13. The method according to claim 9, wherein said second level of results omits B24 through B31.

14. The method according to claim 13, wherein said final level of results comprises:


(B0+B19), (B0+B21), (B1+B20), (B1+B18), (B1+B23), (B2+B20), (B2+B17), (B2+B23), (B3+B16), (B3+B22), (B4+B17), (B4+B18), (B4+B23), (B5+B16), (B5+B22), (B6+B21), (B6+B19), (B7+B20), (B7+B17), (B7+B18), (B8+B20), (B8+B17), (B8+B18), (B9+B21), (B9+B19), (B10+B16), (B10+B22), (B11+B17), (B11+B18), (B11+B23), (B12+B16), (B12+B22), (B13+B20), (B13+B17), (B13+B23), (B14+B20), (B14+B18), (B14+B23), (B15+B21), and (B15+B19).

15. The method according to claim 13, wherein said final level of results consists of:


(B0+B19), (B0+B21), (B1+B20), (B1+B18), (B1+B23), (B2+B20), (B2+B17), (B2+B23), (B3+B16), (B3+B22), (B4+B17), (B4+B18), (B4+B23), (B5+B16), (B5+B22), (B6+B21), (B6+B19), (B7+B20), (B7+B17), (B7+B18), (B8+B20), (B8+B17), (B8+B18), (B9+B21), (B9+B19), (B10+B16), (B10+B22), (B11+B17), (B11+B18), (B11+B23), (B12+B16), (B12+B22), (B13+B20), (B13+B17), (B13+B23), (B14+B20), (B14+B18), (B14+B23), (B15+B21), and (B15+B19).

16. The method according to claim 9, wherein said final level of results omits one or more possible combinations of B0 through B31.

17. The method according to claim 9, wherein said final level of results omits one or more combinations of B0 through B31 that represent invalid combinations of one or more of (C0-CM−1).

18. The method according to claim 9, wherein said final level of results omits one or more combinations of B0 through B31 where the omitted combination(s) would be redundant based on a code specification.

19. The method according to claim 9, wherein said final level of results omits one or more combinations B24 through B31 where the omitted combination(s) would be invalid based on a code specification.

20. The method according to claim 1, further comprising performing an equalization process during one or more of steps (4) and (5).

21. The method according to claim 1, further comprising performing an MLSE process during one or more of steps (4) and (5).

22. The method according to claim 1, further comprising performing an adaptive process during one or more of steps (4) and (5).

23. The method according to claim 1, further comprising performing an adaptive equalization process during one or more of steps (4) and (5).

24. The method according to claim 1, wherein one or more of (C0-CM−1) are constants.

25. The method according to claim 1, wherein one or more of (C0-CM−1) are variable.

26. The method according to claim 1, wherein steps (4) and (5) are implemented in accordance with:




  N
  =
  
    
      
        n
        !
      
      
        
          r
          !
        
        ⁢
        
          
            (
            
              n
              -
              r
            
            )
          
          !
        
      
    
    -
    L
  




wherein:
n represents a number of summer inputs;
r represents a number of summing inputs per kernal; and
L represents a number of invalid combinations.

27. A system for correlating an encoded data word (X0-XM−1) with encoding coefficients (C0-CM−1), wherein each of (X0-XM−1) is represented by one or more bits and each said coefficient is represented by one or more bits, wherein each said coefficient has k possible states, wherein M is greater than 1, comprising:
inputs for each of (X0-XM−1);
a multiplier coupled to each said input;
N summers, each coupled to a different group of outputs of said multipliers, whereby outputs of said N summers form a first layer of correlation results;
one or more additional layers of summers, each said additional layer of summers coupled to outputs of a previous layer of correlation results, said one or more additional layers of summers including a final layer of summers having a final layer of results including a separate correlation output for each possible state of the complete set of coefficients (C0-CM−1);
a magnitude comparator coupled to said final layer of results; and
means for selecting a state of the complete set of coefficients (C0-CM−1) associated with a maximum correlation output.

28. A system for correlating an encoded data word (X0-XM−1) with encoding coefficients (C0-CM−1), wherein each of (X0-XM−1) is represented by one or more bits and each said coefficient is represented by one or more bits, wherein each said coefficient has k possible states, wherein M is greater than 1, comprising:
means for multiplying X0 with states C0(0) through C0(K−1) of said coefficient C0, thereby generating results X0C0(0) through X0C0(K−1);
wherein said means for multiply includes means for multiplying X1-XM−1 with corresponding coefficients (C1-CM−1), respectively;
first means for grouping results of said means for multiplying into N groups and for summing combinations within each of said N groups, thereby generating a first layer of correlation results;
second means for grouping results of said first means for grouping and for summing combinations of results within each group to generate one or more additional layers of results, and repeating this process until a final layer of results includes a correlation output for states of said set of coefficients (C0-CM−1); and
means for comparing magnitudes output of said correlation outputs, thereby identifying a most likely code encoded on said data word.

29. A method for parallel correlation detection, comprising the steps of:
(1) receiving noisy input samples X0, X1, X2, X3, X4, X5, X6, and X7 from which a code is to be extracted;
(2) forming four sets of sample pairs (X0, X1), (X2, X3), (X4, X5), and (X6, X7) from said input samples;
(3) forming four correlation kernels (XiCi+XjCj), (—XiCi+XjCj), (XiCi—XjCj), and (—XiCi—XjCj) for each set of sample pairs formed in step (2), wherein Xi and Xj represent one of the four sample pairs formed in step (2) and wherein Ci and Cj represent predetermined weighting factors;
(4) combining the correlation kernels formed in step (3) to form a fast correlation transform trellis with sixty-four eight-tuple options;
(5) using the sixty-four eight-tuple options formed in step (4) to extract the code from the input samples received in step (1); and
(6) outputting the extracted code.

30. A system for parallel correlation detection, comprising:
a first module that receives noisy input samples X0, X1, X2, X3, X4, X5, X6, and X7 from which a code must be extracted;
a second module that forms four sets of sample pairs (X0, X1), (X2, X3), (X4, X5), and (X6, X7) from said input samples;
a third module that forms four correlation kernels (XiCi+XjCj), (−XiCi+XjCj), (XiCi−XjCj), and (—XiCi—XjCj) for each set of sample pairs formed in said second module, wherein Xi and Xj represent one of the four sample pairs formed in said second module and wherein Ci and Cj represent predetermined weighting factors;
a fourth module that combines the correlation kernels formed in said third module to form a fast correlation transform trellis with sixty-four eight-tuple options; and
a fifth module that compares magnitudes of outputs of said fast correlation transform trellis to identify a most likely code encoded in said noisy input samples.

31. A method for correlating an encoded data word (X0-XM−1) with encoding coefficients (C0-CM−1), wherein each of (X0-XM−1) is represented by one or more bits and each said coefficient is represented by one or more bits, wherein each said coefficient has k possible states, wherein M is greater than 1, comprising:
(1) receiving the encoded data word (X0-XM−1);
(2) multiplying X0 with states of said coefficient C0;
(3) repeating step (2) for data bits (X1-XM−1) and corresponding said coefficients, respectively;
(4) grouping said results of steps (2) and (3) into N groups and summing combinations within each of said N groups, thereby generating a first layer of correlation results;
(5) grouping the results of step (4) and summing combinations of results within each group to generate one or more additional layers of results, and repeating this process until a final layer of results includes a correlation output for each possible state of the set of coefficients;
(6) comparing magnitudes output of said correlation outputs, thereby identifying a most likely code encoded on said data word; and
(7) outputting the most likely code.

32. A method for parallel correlation detection, comprising the steps of:
(1) receiving noisy input samples from which a code must be extracted;
(2) forming at least four sets of sample pairs from said input samples;
(3) forming at least four correlation kernels for each set of sample pairs formed in step (2), wherein Xi and Xj represent one of the sample pairs formed in step (2) and wherein Ci and Cj represent predetermined weighting factors;
(4) combining the correlation kernels formed in step (3) to form a fast correlation transform trellis with at least sixty-four eight-tuple options;
(5) using the at least sixty-four eight-tuple options formed in step (4) to extract the code from the input samples received in step (1); and
(6) outputting the extracted code.