# Patent

## Title
Switch-mode power supply having reduced audible noise

## Abstract

A power supply having an input and an output, includes a power converter coupled between the input and output of the power supply including at least one switch that is controlled by comparing a sensed voltage, the sensed voltage corresponding to a current flowing through the switch, to a reference voltage. A controller, in response to a change detected in a switching frequency of the switch, reduces audible noise generated by the power supply by at least one of: adjusting the reference voltage; adjusting the current sense voltage; or adjusting a resistance used to generate the sensed voltage.

## Background
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention generally relates to the field of power supplies, and more particularly, to reducing audible noise in switch-mode power supplies that have variable switching frequency.
2. Description of the Prior Art
Today, the power supply industry is at the beginning of a major focus shift that puts efficiency improvements across the entire load range in the forefront of customers' performance requirements. This focus on efficiency has been prompted by economic reasons and environmental concerns caused by the continuous growth of the Internet infrastructure and a relatively low energy efficiency of its power delivery system. In fact, the environmental concerns have already prompted introduction of programs and initiatives aimed at reducing the energy waste in power supplies for data-processing applications by challenging power-supply manufacturers to improve efficiency of their products.
A switch-mode power supply comprises a switching power converter for regulating the supply's output voltage. The power converter comprises at least one switch, transformer and a controller that controls the switch according to a switching frequency. This type of power supply suffers from two types of power losses: switching loss and conduction loss. Switching loss is associated with the switching frequency such that this loss increases as switching frequency increases. Conduction loss is associated with current flows in the switch such that the conduction loss increases as the current through the switch increases.
Generally, efficiency across the entire load range requires balancing the switching and conduction losses. This is because efficiency at full load is predominantly determined by conduction losses of semiconductor and magnetic components, whereas efficiency at light/no load is for the most part determined by switching losses of semiconductors and core losses of magnetic components.
One type of switch-mode power supplies operates using a constant switching frequency and another type operates using variable switching frequencies. Regardless of the type, meeting the efficiency requirements at light/no load involves manipulating the switching frequency. In switching power supplies that employ constant switching frequency, cycle skipping (also called burst-mode operation) is employed for meeting the efficiency requirements. In variable switching frequency power supplies, the switching frequency continuously decreases as the load decreases. In some implementations, the burst mode of operation is also used in switch-mode power supplies with variable switching frequency.
However, when the switching frequency is adjusted to be within the upper audible switching frequency range of 200 Hz to 20 kHz, the transformer of the power converter could be excited to generate an unacceptable noise level. As further described below, the switching frequency has an upper audible frequency range and a lower audible frequency range of 20 Hz-200 Hz that is at the lower threshold of the audible range. If the audible noise is above a specified acceptable level, e.g., 25 dB(A)/20.0 uPa for external ac-dc power supplies, appropriate measures should be taken to reduce the audible noise. B. Weisner et al., in “Relevance of magnetostriction and forces for the generation of audible noise of transformer cores,” IEEE Trans. Magnetics, vol. 36, no. 5, September 2000, describe two separate parts for transformer noise that depend on different excitation mechanisms. The first and most dominant part of the transformer noise is caused by magnetization of the core, which is generally assumed to arise from magnetostriction. Under this mechanism, the core dimensions change when subjected to an applied magnetic field. The second part of the transformer noise is caused by electromagnetic forces created by the magnetic field of the currents in the transformer windings. Generally, magnetostriction can cause a mechanical interaction between the core and the windings that leads to a vibration. The mechanical vibration of the transformer is closely related to the magnetic flux swing.
Known methods for reducing the audible noise during burst-mode of operation are disclosed in 1) U.S. Pat. No. 6,477,066 to F. L'Hermite, entitled “Method and apparatus for reducing audible acoustical noise in a power supply transformer by shaping the waveform of a primary side inductor current,” 2) U.S. Pat. No. 6,525,514 to B. Balakrishnan, et all, entitled “Method and apparatus for reducing audio noise in a switching regulator,” and 3) U.S. Patent Application Pub. No. 2006/0171179 by J. W. Hall and C. Basso, entitled “Low audible noise power supply method and controller therefore.” U.S. Pat. No. 6,477,066 and U.S. Patent Application Pub. No. 2006/0171179 disclose reducing the audible noise in burst-mode of operation by shaping the envelope of the switch current pulses. U.S. Pat. No. 6,525,514 discloses reducing the audio noise by presetting the switching frequency values above the audible range and by decreasing the switch current limit as the load decreases until the lowest current limit has been reached. This limit is low enough that the flux density in the core of the transformer does not produce unacceptable audible noise. Specifically, in U.S. Pat. No. 6,525,514, constant switching frequency power converters with on/off feedback control and with a relatively complex cycle skipping algorithm are considered.
While the above methods apply to burst mode operation, in switch-mode power supplies with variable switching frequency, the audible noise can be completely eliminated by preventing the switching frequency from dropping below the upper threshold of the audible range. This can be achieved directly by limiting the maximum switching period or by limiting the maximum off time. The same can be achieved indirectly by monitoring the switching frequency to detect when the switching frequency drops close to the upper threshold of the audible range and, then, instantly increasing the switching frequency, as described in U.S. Pat. No. 7,202,609 to W. H. M Langeslag and J. W. Strijker, entitled “Noise reduction in a power converter.” It should be noted that U.S. Pat. No. 7,202,609 is directed towards the problem of reducing the audible noise in power converters with variable switching frequency. The disclosed arrangement in this patent cover a power supply design that decreases the switching frequency as the load increases. Thus, the switching frequency can only drop below the upper threshold of the audible range at high load conditions that are associated with when exceptional operating circumstances exist, such as a short circuit, initial power up, or sudden turn off situations. U.S. Pat. No. 7,202,609 is not directed to audible noise reduction in switch-mode power supplies under normal (non-exceptional) operating conditions where the switching frequency continuously decreases as the load decreases.
Therefore, there exists a need for a system and method for audible noise reduction in switch-mode power supplies with variable switching frequency where the switching frequency continuously decreases as the load decreases.
SUMMARY OF THE INVENTION
Briefly, according to the present invention, a power supply that provides output power to a load comprises a power converter coupled to the load. The power converter has at least one switch that is operated at a switching frequency that results in switch current flows (or pulses) through the switch. The switching frequency has an audible frequency range that results in unacceptable audible noise that is reduced by the present invention. A controller regulates the output power by controlling the switching frequency based on a feedback signal that is coupled to the load. The controller comprises a comparator and a current sense resistor for comparing a reference voltage with a current sense voltage developed across the current sense resistor by the switch current flows. An audible noise reduction circuit adjusts either one of the reference voltage, the current sense voltage or the current sense resistor based on a detected switching frequency such that the controller maintains the switching frequency above the audible frequency range in response to an adjustment. The switching frequency can be detected based on either one of a switch drive signal, the feedback signal or a current flow signal through the load. The audible noise reduction circuit can be disabled when the detected switching frequency is within a lower audible frequency range, e.g., 20 Hz-200 Hz, that is at the lower threshold of the audible frequency range.



BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 shows the block diagram of a conventional switch-mode power supply.
FIG. 2 shows the relationship between the switching frequency and load current of a converter in the power supply of FIG. 1.
FIGS. 3 (a) and (b) show various operating waveforms of the power supply of FIG. 1.
FIG. 4 shows the block diagram of a power supply according to one embodiment of the invention.
FIG. 5 shows the relationship between the switching frequency and load current of a converter in the power supply of FIG. 4.
FIGS. 6 (a) and (b) show various operating waveforms of the power supply of the invention according to one audible noise reduction embodiment.
FIGS. 7 (a) and b) show various operating waveforms of the power supply of the invention according to another audible noise reduction embodiment.
FIGS. 8 (a) and b) show various operating waveforms of the power supply of the invention according to yet another audible noise reduction embodiment.
FIG. 9 shows additional relationships between the switching frequency and load current according to other implementations of the invention.
FIG. 10 shows yet another method of the present invention for the audible noise reduction.
FIG. 11 shows the block diagram of a power supply according to another embodiment of the invention.
FIG. 12 shows the block diagram of a power supply according to yet another embodiment of the invention.
FIG. 13 shows one implementation according to the embodiment of the present invention presented in FIG. 4.
FIG. 14 shows waveforms that illustrate the operation of the circuit in FIG. 13 around ILoad=ILoad1 shown in FIG. 10, when the switching frequency decreases.
FIG. 15 shows waveforms that illustrate the operation of the circuit in FIG. 13 around ILoad=ILoad2 shown in FIG. 10, when the switching frequency decreases.
FIG. 16 shows waveforms that illustrate the operation of the circuit in FIG. 13 around ILoad=ILoad3 shown in FIG. 10, when the switching frequency decreases.
FIG. 17 shows waveforms that illustrate the operation of the circuit in FIG. 13 around ILoad=ILoad3 shown in FIG. 10, when the switching frequency increases.
FIG. 18 shows waveforms that illustrate the operation of the circuit in FIG. 13 around ILoad=ILoad2 shown in FIG. 10, when the switching frequency increases.
FIG. 19 shows waveforms that illustrate the operation of the circuit in FIG. 13 around ILoad=ILoad1 shown in FIG. 10, when the switching frequency increases.
FIG. 20 shows one circuit embodiment for decreasing the peak value of the main switch current pulses.
FIG. 21 shows another circuit embodiment for decreasing the peak value of the main switch current pulses.
FIG. 22 shows yet another circuit embodiment for decreasing the peak value of the main switch current pulses.
FIG. 23 shows one implementation according to the embodiment of the present invention presented in FIG. 11.
FIG. 24 shows principle of operation of the circuit in FIG. 23.
FIG. 25 shows the method of the present invention for the audible noise reduction presented in FIG. 12 when the switching frequency is monitored indirectly by monitoring the feedback voltage.



DETAILED DESCRIPTIONS OF THE PREFERRED EMBODIMENTS
The present invention reduces audible noise in a switch-mode power supply that decreases the switching frequency continuously as the load decreases. The power supply of the invention has a power converter coupled between an input voltage source and a load. The power converter has one or more switches that are switched to on and off states with a variable switching frequency according to load conditions to provide regulated output power to the load. The current through each one of the one or more switches, when turned on, rises in accordance with a switching duty cycle to form switch current flows or pulses. Such current flows or pulses have peak values that are reached before a switch is turned off.
In one embodiment, the audible noise is reduced by causing the peak value of the switch current pulses to decrease in discrete steps so that the switching frequency is increased above the upper threshold of audible range in order to maintain regulated power at the load as the load decreases. The peak value of the current pulses is reduced to a level that the corresponding magnetic flux swing in the core of the transformer of the power converter does not produce unacceptable audible noise. Various methods are disclosed for detecting the switching frequency and for decreasing the peak value of the switch current pulses for keeping the switching frequency above the audible noise range. According to one embodiment of the invention, the switching frequency is directly monitored to detect whether it has dropped close to the upper threshold of the audible range fswThH1. According to another embodiment, the switching frequency is indirectly monitored to detect whether it has dropped close to the upper threshold of the audible range fswThH1. Such indirect monitoring of the switching frequency can be accomplished either by monitoring the load current ILoad or by monitoring the feedback voltage VFB.
When the peak value of the main switch current pulses is decreased, according to any one the embodiments described above, the switching frequency is increased in order to provide the same power to the load. However, at very light loads and at no load, an increased switching frequency results in increased switching losses. In order to meet efficiency requirements that limit the maximum input power at very light loads and at no load, the audible noise reduction circuit is disabled when the switching frequency is so low, e.g., 20 Hz-200 Hz, that a magnetic flux swing in the core of the transformer does not produce unacceptable audible noise (due to the nonlinear sensitivity of the human ear to the audible frequencies). This switching frequency range is herein called a lower audible frequency range which is at the lower threshold of the audible frequency range. In such situation, the switching frequency is allowed to decrease to the lower audible switching frequency range by increasing the peak value of the switch current pulses to their original peak value.
Before proceeding with describing the present invention, a brief description of the prior art is provided in connection with FIG. 1, which shows the block diagram of a known switch-mode power supply, which in this example, operates in a discontinuous conduction mode (DCM). The power supply comprises a power converter having a variable switching frequency fsw that is coupled between an input voltage source, e.g. an AC line, and a load. For example, the load can correspond to the load of a computing device that varies based on corresponding operating conditions. The power converter of the power supply of FIG. 1 is assumed to be a flyback converter having a main switch that operates according to a variable switching frequency. The current flowing through a converter switch isw comprises current pulses having a peak value IPeak. As stated above, if the peak value IPeak is not sufficiently low, the corresponding magnetic flux swing in the core of the transformer of the flyback converter (not shown) can produce unacceptable audible noise.
The switch-mode power supply of FIG. 1 also includes a known controller forming a continuous feedback circuit that couples the output of the power supply to the power converter. The controller varies the switching-frequency fsw based on a controller input voltage that corresponds to the conditions of the load. The controller input voltage is applied to an error amplifier (EA) having an input reference voltage Vout, Ref. A voltage-controlled oscillator (VCO) is controlled by a feedback voltage VFB at the output of the EA. The output of the VCO Vvco is coupled to a “Set” input S of a flip-flop FF. The VCO is configured to be responsive to the feedback voltage VFB, which decreases as the load decreases in order to decrease the switching frequency fsw. In this way, the VCO controls the converter's main switches with a variable switching frequency fsw. A comparator has one input coupled to a current-sense (CS) voltage Vcs, which is developed across a current-sense resistor Rcs based on sensed switch current isw. Another input of the comparator is coupled to a current-sense reference voltage Vcs, Ref. The output of the comparator is coupled to a “Reset” input R of the flip-flop FF so that a duty cycle initiated by the VCO is terminated by the comparator when the current-sense voltage VCS reaches the reference voltage level VCS,Ref. In this way, the output of the comparator is used for resetting a duty cycle initiated by the VCO. The FF also outputs a gate drive voltage VGD that controls turning the converter's main switch on and off.
FIG. 2 shows the relationship between the switching frequency fsw and load current ILoad of the power supply of FIG. 1. For simplicity, in FIG. 2, a linear relationship between the switching frequency and load current is assumed, which is a good approximation for a DCM operating converter such as forward, half-bridge, full-bridge, flyback, boost, etc. As shown in FIG. 2, the switching frequency fsw has an audible range within the range of fAudibleL and fAudibleH. FIG. 2 shows that the switching frequency fsw enters the audible range at load current ILoad1. The minimum switching frequency, fswMin, is obtained at no load, which is typically above the lower threshold of the audible range, fAudibleL.
FIGS. 3 (a) and (b) show graphs of various waveforms of the power supply of FIG. 1 described. FIG. 3(a) shows the waveforms at a first switching frequency and FIG. 3(b) shows that waveforms at a second switching frequency such that the first switching frequency is twice the second switching frequency. FIGS. 3(a) and (b) show that under the prior art method, if the peak value of the switch current pulses IPeak is to remain constant, a linear relationship should exist between the switching frequency fsw and load current ILoad so that when the second switching frequency fsw is decreased to be one half of the first switching frequency, as illustrated in FIG. 3(b), the average switch current isw and consequently the load current ILoad is also decreased by the same ratio, e.g., one half.
FIG. 4 shows the block diagram of a power supply according to one embodiment of the invention having a power converter that can be arranged to any suitable configuration, including but not limited to forward, half-bridge, full-bridge, flyback, or boost configurations. The power converter supplies regulated power to the load from an input power source. The power converter has one or more switches that are switched to on and off states with a variable switching frequency according to operation conditions of the load via a feedback controller circuit. The current through each one of the one or more switches when turned on rises in accordance with a switching duty cycle to form switch current pulses having peak values that are reached before a switch is turned off in response to a corresponding switch driving signal provided by the controller via a Switching Device Timing & Drive (SDTD) block. The SDTD block is responsive to the output of a VCO and a comparator. As described in connection with FIG. 1, the output of the VCO is controlled by the feedback voltage VFB at the output of the EA and the output of the comparator is used for resetting a duty cycle initiated by the VCO when the current-sense voltage VCS reaches reference voltage level VCS,Ref.
An audible noise reduction (ANR) circuit is coupled to the controller for reducing the audible noise according to various embodiments of the invention. More specifically, the embodiment shown in FIG. 4 reduces audible noise by directly monitoring the switching frequency by monitoring the switch driving signal. The ANR circuit comprises a first block for the detection of the switching frequency at the upper threshold of the audible range. A second block programs the peak value of the main switch current pulses and a third block disables the audible noise reduction circuit at very light loads and at no load. As shown, the inputs of the first block and the third block are coupled to the switch driving signal allowing for direct monitoring of the switching frequency. In order to reduce the peak value of the switch current pulses, the output of the peak-current programming block modifies either the reference voltage value of the current-sense voltage VCS,Ref, the current-sense voltage VCS itself, or the value of the current-sense resistor RCS.
FIG. 5 shows a graph of the relationship between the load current ILoad and the switching frequency fsw according to one embodiment of the present invention where audible noise is reduced by controlling the switching frequency so that it stays above the audible range of fAudibleL to fAudibleH as the load decreases, within a high and low threshold switching frequency range of fswThH1 and fswThH2. The load currents ILoad1 and ILoad2 correspond to decreasing load conditions of the power supply operating according to this embodiment of the invention. In this embodiment, the power supply is operated to decrease the peak value of the switch current pulses IPeak in discrete steps from IPeak1 to IPeak2 to IPeak3 until the peak value of the switch current pulses is at a level which is sufficiently low such that the generated switch current pulses do not produce magnetic flux swing in the core of the transformer that causes unacceptable audible noise.
In an embodiment, after detecting that the switching frequency has dropped close to the upper threshold of the audible range, the peak value of the main switch current pulses is decreased in discrete steps. Consequently, the switching frequency increases by Δfsw, as shown in FIG. 5, in order to provide the same power to the load.
FIGS. 6(a) and 6(b) show comparative waveforms of one embodiment where the peak value of the main switch current pulses is decreased by ΔIpeak by decreasing the reference value of the current-sense voltage, VCS,Ref. FIGS. 7(a) and 7(b) show comparative waveforms according to another embodiment where the peak value of the main switch current pulses is decreased by ΔIpeak by adding a DC bias Vcs,bias to the current-sense voltage. FIGS. 8(a) and 8(b) show comparative graphs according to yet another embodiment where the peak value of the switch current pulses is decreased by ΔIpeak by increasing the value of the current-sense resistor from RCS1 to RCS2.
FIG. 9 shows a graph of load current ILoad relative to the switching frequency fsw under an embodiment where a hysteresis is added to the control of the switching frequency in order to prevent the oscillation of the switching frequency when the peak value of the main switch current pulses changes between two consecutive discrete values. It should be noted that the switching frequency control by employing three discrete peak values of the main switch current pulses as shown in FIGS. 5 and 9 is an illustrative example and that the number of the discrete peak values of the main switch current pulses can be selected from a minimum value of two to a finite large number.
At very light loads and at no load, an increased switching frequency results in increased switching losses. Consequently, meeting standard requirements that limit the maximum input power at very light loads and at no load can be an issue. FIG. 10 shows a graph of load current ILoad relative to the switching frequency fsw at very light loads and at no load, where the audible noise reduction circuit is disabled and the switching frequency is decreased to its original value at ILoad=ILoad3. In fact, at ILoad=ILoad3, the switching frequency is decreased by increasing the peak value of the main switch current pulses to the original peak value IPeak1. It should be noted that at very light loads and at no load, the original switching frequency is, typically, close to the lower threshold of the audible range. Therefore, a larger peak value of the main switch current pulses and, consequently, a larger magnetic flux swing in the core of the transformer will not produce unacceptable audible noise as a result of the nonlinear sensitivity of the human ear to the audible frequencies.
FIG. 11 shows another embodiment of the present invention that reduces audible noise by monitoring the load current, which indirectly corresponds to the switching frequency of the power converter. As shown, a signal representing the load current is applied to the inputs of the first block and the third block of the ANR circuit. FIG. 12 shows yet another embodiment of the present invention that reduces audible noise by monitoring the feedback voltage, which also indirectly corresponds to the switching frequency. As shown, the feedback voltage VFB at the output of the EA and input of the VCO is applied to the inputs of the first block and the third block of the ANR circuit.
FIG. 13 shows an example of the implementation of the ANR circuit. Key waveforms that illustrate the operation of the ANR circuit in FIG. 13 are presented in FIGS. 14-19. Specifically, key waveforms at ILoad=ILoad1, ILoad=ILoad2, and ILoad=ILoad3, when the load current decreases, are presented in FIGS. 14-16, respectively; whereas, key waveforms around ILoad=ILoad3, ILoad=ILad2, and ILoad=ILoad1, when the load current increases, are presented in FIGS. 17-19, respectively. Block H in FIG. 13 implements the control of the switching frequency at the upper threshold of the audible range around load currents ILoad=ILoad1 and ILoad=ILoad2; whereas, Block L in FIG. 13 implements the control of the switching frequency inside the audible range at very light loads around ILoad=ILoad3. The output signals OUT1 and OUT2 of the control circuit in FIG. 13 control the peak value of the main switch current pulses as defined in Table I.







TABLE I







Relationship between peak value of the main switch


current pulses and control signals OUT1 and OUT2











OUT1
OUT2
IPeak







0
0
Ipeak1



1
0
IPeak2



1
1
IPeak3










In FIG. 13, the switching frequency, i.e., the switching period is monitored by converting time to voltage. The ramp voltages, vRampH and vRampL, are generated by charging the respective ramp capacitors CH and CL with a constant current, IH and IL, respectively. At the beginning of a switching period, the ramp capacitors are discharged by briefly turning on the respective ramp switches SWH1 and SWL1 in parallel with the ramp capacitors. The control signal for the ramp switches is obtained by differentiating the gate drive signal of the main switch.
The detection of the switching frequency at the upper threshold of the audible range is achieved by comparing ramp voltage vRampH at the noninverting input of comparators COMPH1 and COMPH2 to the corresponding reference voltage, VCompH1—Ref and VCompH2—Ref, at the inverting input of comparators COMPH1 and COMPH2. The detection of the switching frequency inside the audible range at very light loads is achieved by comparing ramp voltage vRampL at the noninverting input of comparator COMPL to the reference voltage VcompL—Ref at the inverting input of comparator COMPL. To control the switching frequency according to the method shown in FIG. 10, each reference voltage has two discrete levels. For example, the reference voltage of comparator COMPH1 has a higher level equal to the voltage of reference voltage source VRefH as long as output Q of D flip-flop FFH1 is at LOW level. When output Q of D flip-flop FFH1 becomes HIGH, switch SWH2 turns on, and the reference voltage of comparator COMPH1 takes a lower level. The lower level is determined by the voltage divider RH1-RH2. The reference voltage of comparators COMPH2 and COMPL is determined similarly to the reference voltage of comparator COMPH1. However, it should be noted that the reference voltage of comparator COMPL is controlled by inverting output Q of D flip-flop FFL. It should also be noted that the ramp voltages are clamped by Zener diodes ZDH and ZDL to a level slightly higher than the voltage of the respective reference voltage sources.
The output voltage of a comparator is at HIGH level if the switching frequency is equal or smaller than the relevant reference frequency, i.e., if the switching period is equal or greater than the relevant reference period. The information about the switching frequency obtained in the current switching period is used in the next switching period to control the peak value of the main switch current pulses. In fact, the voltage level at the output of a comparator is stored in a corresponding D flip-flop at the positive edge of the gate drive signal. It should be noted that the output voltage of a comparator can easily meet the setup-time requirement of a positive-edge triggered D flip-flop due to the delay time of the comparator. Initially, all D flip-flops are reset through the direct reset (DR) pins.
The waveforms in FIG. 14 illustrate the operation of the circuit in FIG. 13 at ILoad=ILoad1 shown in FIG. 10. At ILoad=ILoad1, the switching frequency decreases close to the upper threshold of the audible range. Consequently, the peak value of the main switch current pulses decreases from Ipeak1 to IPeak2, resulting in an increased switching frequency Δfsw shown in FIG. 10.
Before instant t=T0, the output voltage of all three D flip-flops, FFH1, FFH2, and FFL, is LOW and, therefore, both output signals OUT1 and OUT2 are LOW, and IPeak=IPeak1. The reference voltage of comparators COMPH1 and COMPH2, VCompH1—Ref and VCompH2—Ref, has HIGH value, whereas, the reference voltage of comparator COMPL, VCompL—Ref, has LOW value. During switching cycle [T0, T1], the switching frequency decreases below threshold frequency fswThH1, i.e., the switching period increases above threshold period TswThH1, and ramp voltage vRampH crosses the reference voltage level of comparators COMPH1 and COMPH2. Consequently, the output voltage of comparators COMPH1 and COMPH2 becomes HIGH. The D input of D flip-flop FFH1 also becomes HIGH through the OR gate ORH1; however, the D input of D flip-flop FFH2 stays LOW because it is inhibited by the LOW output voltage of D flip-flop FFH1 through AND gate ANDH1. At the beginning of the next switching cycle, [T1, T2], at the positive edge of the gate drive signal, the data at the D input of the D flip-flops is transferred to the Q output of the flip-flops, resulting in QFFH1=1, QFFH2=0, and QFFL=0, and, consequently, OUT1=1 and OUT2=0. According to Table I, the peak value of the main switch current pulses decreases from IPeak1 to IPeak2. It should be noted in FIG. 14 that the gate drive pulse width in switching cycle [T0, T1] is larger than the gate drive pulse width in the following switching cycles. To keep the output voltage of the power converter constant, the feedback voltage will increase, resulting in an increased switching frequency.
When output Q of D flip-flop FFH1 becomes HIGH, the reference voltage of comparator COMPH1 decreases from HIGH to LOW level. As long as the switching frequency is lower than fswThH2, i.e., the switching period is larger than TswThH2, ramp voltage vRampH will always cross the LOW reference voltage level of comparator COMPH1 and, therefore, output Q of D flip-flop FFH1 will stay HIGH.
The waveforms in FIG. 15 illustrate the operation of the circuit in FIG. 13 at ILoad=ILoad2 shown in FIG. 10, where the switching frequency decreases the second time close to the upper threshold of the audible range and, consequently, the peak value of the switch current pulses decreases from IPeak2 to IPeak3, resulting in an increased switching frequency Δfsw shown in FIG. 10.
Before instant t=T0, the output voltage of D flip-flop FFH1 is HIGH and the output voltage of D flip-flops FFH2, and FFL1 is LOW. Therefore, the signals that control the peak value of the main switch current pulses are OUT1=1 and OUT2=0, and IPeak=IPeak2. The reference voltage of comparators COMPH1 and COMPH2, VCompH1—Ref and VcompH2—Ref, is LOW and HIGH, respectively. The reference voltage of comparator COMPL, VCompL—Ref, is also LOW. During switching cycle [T0, T1], the switching frequency decreases below threshold frequency fswThH1, i.e., the switching period increases above threshold period TswThH1, and ramp voltage vRampH crosses the reference voltage level of comparator COMPH2. Consequently, the output voltage of comparator COMPH2 becomes HIGH and the D input of D flip-flop FFH2 also becomes HIGH through the AND gate ANDH1 because now it is inhibited by the HIGH output voltage of D flip-flop FFH1. At the beginning of the next switching cycle, [T1, T2], at the positive edge of the gate drive signal, the HIGH logic level at the D input of D flip-flop FFH2 is transferred to its Q output, resulting in QFFH2=1. The output of the other two D flip-flops is QFFH1=1 and QFFL=0. Consequently, both output signals become HIGH, i.e., OUT1=1 and OUT2=1. According to Table I, the peak value of the main switch current pulses decreases from IPeak2 to IPeak3. It should be noted in FIG. 15 that the gate drive pulse width in switching cycle [T0, T1] is larger than the gate drive pulse width in the following switching cycles. To keep the output voltage of the power converter constant, the feedback voltage will increase, resulting in an increased switching frequency.
When the output voltage of D flip-flop FFH2 becomes HIGH, the reference voltage of comparator COMPH2 decreases from HIGH to LOW level. As long as the switching frequency is lower than fswThH2, i.e., the switching period is larger than TswThH2, ramp voltage vRampH will always cross the LOW reference voltage level of comparator COMPH2 and, therefore, the output voltage of D flip-flop FFH2 will stay HIGH. It should be noted in FIG. 15 that the HIGH output voltage of D flip-flop FFH2 will force the D input of D flip-flop FFH1 to HIGH logic level through the OR gate ORH1.
The waveforms in FIG. 16 illustrate the operation of the circuit in FIG. 13 at very light load ILoad=ILoad3 shown in FIG. 10. At ILoad=ILoad3, the switching frequency decreases to threshold level fswThL1 inside the audible range. Consequently, the peak value of the switch current pulses increases from IPeak3 to IPeak1, resulting in a further decreased switching frequency as shown in FIG. 10.
Before instant t=T0, the output voltage of D flip-flops FFH1 and FFH2 is HIGH and the output voltage of D flip-flop FFL is LOW. Therefore, both output signals are HIGH, i.e., OUT1=1 and OUT2=1, and Ipeak=IPeak3. The reference voltage of all three comparators COMPH1, COMPH2, and COMPL, VCompH1—Ref, VCompH2—Ref, and VCompL—Ref, is LOW. During switching cycle [T0, T1], the switching frequency decreases below threshold frequency fswThL1, i.e., the switching period increases above threshold period TswThL1, and ramp voltage vRampL crosses the reference voltage level of comparator COMPL. Consequently, the output voltage of comparator COMPL and the D input of D flip-flop FFL become HIGH. At the beginning of the next switching cycle, [T1, T2], at the positive edge of the gate drive signal, the HIGH logic level at the D input of D flip-flop FFL is transferred to its Q output, resulting in QFFL=1. Consequently, through the AND gates ANDH2 and ANDH3, both output signals become LOW, i.e., OUT1=0 and OUT2=0. According to Table I, the peak value of the main switch current pulses increases from IPeak3 to IPeak1. It should be noted in FIG. 16 that the gate drive pulse width in switching cycle [T0, T1] is smaller than the gate drive pulse width in the following switching cycles. To keep the output voltage of the power converter constant, the feedback voltage will decrease, resulting in a further decreased switching frequency.
When output Q of D flip-flop FFL becomes HIGH, the reference voltage of comparator COMPL increases from LOW to HIGH level. It should be noted that because the gate drive pulse width is much smaller than the switching period and the output voltage regulation loop is fast, during switching period [T1, T2] the feedback voltage will decrease to the lower level well before the end of the switching cycle. As long as the switching frequency is lower than fswThL2, i.e., the switching period is larger than TswThL2, ramp voltage vRampL will always cross the HIGH reference voltage level of comparator COMPL and, therefore, output Q of D flip-flop FFL will stay HIGH, keeping both output signals that control the peak value of the main switch current pulses LOW, i.e., OUT1=0 and OUT2=0.
FIGS. 17-19 illustrate key waveforms of the operation of the circuit in FIG. 13 around ILoad=ILoad3, ILoad=ILoad2, and ILoad=ILoad1, when the load current increases, respectively. It can be seen in FIGS. 17-19 that the operation of the circuit in FIG. 13 is similar to the operation when the load current decreases, just in the opposite direction.
FIG. 20 shows one implementation of the peak-current programming block in FIGS. 4, 11 and 12 according to Table I and according to FIG. 6, where the peak value of the switch current pulses is decreased by decreasing the reference value of the current-sense voltage, VCS,Ref. The peak-current programming block includes two switches SW1 and SW2 with respective control signals OUT1 and OUT2. The control signals correspond to the outputs of the first and third block of the ANR circuit. The output VCS,Ref of the peak-current programming block is directly coupled to the comparator's input of the controller. Current-sense reference voltage VCS,Ref(IPeak1) corresponds to the maximum peak value of the main switch current pulses, IPeak1. When both control signals are LOW, i.e., OUT1=0 and OUT2=0, both switches SW1 and SW2 are open, resulting in

VCS,Ref=VCS,Ref(IPeak1).  (1)

When control signals are OUT1=1 and OUT2=0, the current-sense reference voltage is determined as





  
    
      
        
          
            V
            
              CS
              ,
              Ref
            
          
          =
          
            
              
                R
                1
              
              
                
                  R
                  1
                
                +
                
                  R
                  3
                
              
            
            ·
            
              
                V
                
                  CS
                  ,
                  Ref
                
              
              ⁡
              
                (
                
                  I
                  
                    Peak
                    ⁢
                    
                      
                    
                    ⁢
                    1
                  
                
                )
              
            
          
        
        ,
      
    
    
      
        (
        2
        )
      
    
  




which corresponds to the second peak value of the main switch current pulses, IPeak2. Finally, when both control signals are HIGH, i.e., OUT1=1 and OUT2=1, the current-sense reference voltage is determined as





  
    
      
        
          
            V
            
              CS
              ,
              Ref
            
          
          =
          
            
              
                
                  R
                  1
                
                ||
                
                  R
                  2
                
              
              
                
                  R
                  1
                
                ||
                
                  
                    R
                    2
                  
                  +
                  
                    R
                    3
                  
                
              
            
            ·
            
              
                V
                
                  CS
                  ,
                  Ref
                
              
              ⁡
              
                (
                
                  I
                  
                    Peak
                    ⁢
                    
                      
                    
                    ⁢
                    1
                  
                
                )
              
            
          
        
        ,
      
    
    
      
        (
        3
        )
      
    
  




which corresponds to the minimum peak value of the main switch current pulses, IPeak3.

FIG. 21 shows another implementation of the peak-current programming block in FIGS. 4, 11 and 12 according to Table I and according to FIG. 7, where the peak value of the main switch current pulses is decreased by adding a dc bias to the current-sense voltage, vCS. The peak-current programming block receives control signals from the first and third block of the ANR circuit. The output vCS of the peak-current programming block is coupled to the current-sense voltage input, vCS, of the comparator of the controller. When both control signals are LOW, i.e., OUT1=0 and OUT2=0, both switches SW1 and SW2 are open and the current-sense voltage vCS is equal to the voltage across the current-sense resistor. It should be noted in FIG. 21 that resistor R3 and capacitor C1 make the conventional low-pass filter at the current-sense input of a control circuit. When control signals are OUT1=1 and OUT2=0, the current-sense voltage is determined as




  
    
      
        
          V
          CS
        
        =
        
          
            
              
                
                  R
                  1
                
                
                  
                    R
                    1
                  
                  +
                  
                    R
                    3
                  
                
              
              ·
              
                R
                CS
              
            
            ⁢
            
              I
              SW
            
          
          +
          
            
              
                R
                3
              
              
                
                  R
                  1
                
                +
                
                  R
                  3
                
              
            
            ·
            
              
                V
                CC
              
              .
            
          
        
      
    
    
      
        (
        4
        )
      
    
  




Eq. (4) can be rewritten as





  
    
      
        
          
            V
            CS
          
          =
          
            
              
                
                  R
                  1
                
                
                  
                    R
                    1
                  
                  +
                  
                    R
                    3
                  
                
              
              ·
              
                (
                
                  
                    
                      R
                      CS
                    
                    ⁢
                    
                      I
                      SW
                    
                  
                  +
                  
                    
                      
                        R
                        
                          3
                          ⁢
                          
                            
                          
                        
                      
                      
                        R
                        1
                      
                    
                    ·
                    
                      V
                      CC
                    
                  
                
                )
              
            
            ≈
            
              
                
                  R
                  CS
                
                ⁢
                
                  I
                  SW
                
              
              +
              
                
                  
                    R
                    3
                  
                  
                    R
                    1
                  
                
                ·
                
                  V
                  CC
                
              
            
          
        
        ,
      
    
    
      
        (
        5
        )
      
    
  




because R3<<R1. The second term in Eq. (5) represents the dc bias of the current-sense voltage. Finally, when both control signals are HIGH, i.e., OUT1=1 and OUT2=1, the current-sense voltage is determined as





  
    
      
        
          V
          CS
        
        =
        
          
            
              
                
                  R
                  1
                
                ||
                
                  R
                  2
                
              
              
                
                  R
                  1
                
                ||
                
                  
                    R
                    2
                  
                  +
                  
                    R
                    3
                  
                
              
            
            ·
            
              (
              
                
                  
                    R
                    CS
                  
                  ⁢
                  
                    I
                    SW
                  
                
                +
                
                  
                    
                      R
                      3
                    
                    
                      
                        R
                        1
                      
                      ||
                      
                        R
                        2
                      
                    
                  
                  ·
                  
                    V
                    CC
                  
                
              
              )
            
          
          ≈
          
            
              
                R
                CS
              
              ⁢
              
                I
                SW
              
            
            +
            
              
                
                  R
                  3
                
                
                  
                    R
                    1
                  
                  ||
                  
                    R
                    2
                  
                
              
              ·
              
                
                  V
                  CC
                
                .
              
            
          
        
      
    
    
      
        (
        6
        )
      
    
  




It should be noted that the dc bias of the current-sense voltage in Eq. (6) is larger than that in Eq. (5).

FIG. 22 shows yet another implementation of the peak-current programming block in FIGS. 4, 11 and 12 according to Table I and according to FIG. 8, where the peak value of the main switch current pulses is decreased by effectively increasing the value of the current-sense resistor. The peak-current programming block receives control signals from the first and third block of the ANR circuit. The output vCS of the peak-current programming block is coupled to the current-sense voltage input, vCS, of the comparator of the controller. When both control signals are LOW (OUT1=0 and OUT2=0), which corresponds to the maximum peak value of the switch current pulses, IPeak1, both switches SW1 and SW2 are closed and, therefore, current-sense voltage




  
    
      
        
          V
          CS
        
        =
        
          
            
              
                
                  R
                  1
                
                ||
                
                  R
                  2
                
              
              
                
                  R
                  1
                
                ||
                
                  
                    R
                    2
                  
                  +
                  
                    R
                    3
                  
                
              
            
            ·
            
              I
              SW
            
          
          ⁢
          
            R
            CS
          
        
      
    
    
      
        (
        7
        )
      
    
  




is smaller than the voltage across the current-sense resistor. The effective current-sense resistor in Eq. (7) is determined as





  
    
      
        
          
            R
            
              CS
              ,
              eff
            
          
          ⁡
          
            (
            
              I
              
                Peak
                ⁢
                
                  
                
                ⁢
                1
              
            
            )
          
        
        =
        
          
            
              
                R
                1
              
              ||
              
                R
                2
              
            
            
              
                R
                1
              
              ||
              
                
                  R
                  2
                
                +
                
                  R
                  3
                
              
            
          
          ·
          
            
              R
              CS
            
            .
          
        
      
    
    
      
        (
        8
        )
      
    
  




When the control signals are OUT1=1 and OUT2=0, which corresponds to the second peak value of the main switch current pulses, IPeak2, only SW2 is closed and, therefore, current-sense voltage is obtained as





  
    
      
        
          V
          CS
        
        =
        
          
            
              
                R
                2
              
              
                
                  R
                  2
                
                +
                
                  R
                  3
                
              
            
            ·
            
              I
              SW
            
          
          ⁢
          
            
              R
              CS
            
            .
          
        
      
    
    
      
        (
        9
        )
      
    
  




The effective current-sense resistor in Eq. (9) is determined as





  
    
      
        
          
            R
            
              CS
              ,
              eff
            
          
          ⁡
          
            (
            
              I
              
                Peak
                ⁢
                
                  
                
                ⁢
                2
              
            
            )
          
        
        =
        
          
            
              R
              2
            
            
              
                R
                2
              
              +
              
                R
                3
              
            
          
          ·
          
            
              R
              CS
            
            .
          
        
      
    
    
      
        (
        10
        )
      
    
  




The effective current-sense resistor in Eq. (10) is larger than that in Eq. (8). Finally, when both control signals are HIGH (OUT1=1 and OUT2=1), which corresponds to the minimum peak value of the main switch current pulses, IPeak3, both switches SW1 and SW2 are open and, therefore, the current-sense voltage is equal to the voltage across the current-sense resistor, i.e.,

VCS=ISWRCS.  (11)

It follows from Eqs. (11), (10), and (8) that

RCS,eff(IPeak3)=RCS>RCSRCS,eff(IPeak2)>RCS,eff(IPeak1).  (12)

FIG. 23 shows one implementation according to the embodiment presented in FIG. 11, where the switching frequency is indirectly monitored by monitoring the load current. Principle of operation of the circuit in FIG. 23 is presented in FIG. 24.
The output signals OUT1 and OUT2 of the circuit in FIG. 23 control the peak value of the main switch current pulses as defined in Table I.
As long as the load current is larger than ILoad1 (shown in FIG. 10), the output voltage of all three comparators is LOW and the light-emitting diodes (LEDs) of opto-couplers OC1 and OC2 conduct. Therefore, both output signals are LOW, i.e., OUT1=0 and OUT2=0, and IPeak=IPeak1. When the load current decreases to ILoad1, the output voltage of comparator COMP1 becomes HIGH, the LED of opto-coupler OC1 stops to conduct, resulting in OUT1=1. With OUT1=1 and OUT2=0, the peak value of the main switch current pulses decreases from IPeak1 to IPeak2. When the load current decreases to ILoad2 (shown in FIG. 10), the output voltage of comparator COMP2 also becomes HIGH, the LED of opto-coupler OC2 stops to conduct, resulting in OUT2=1. With OUT1=1 and OUT2=1, the peak value of the main switch current pulses decreases from IPeak2 to IPeak3. When the load current decreases to ILoad3 (shown in FIG. 10), the output voltage of comparator COMP3 becomes HIGH, switches Q1 and Q2 conduct and pull down the output of comparators COMP1 and COMP2 to LOW level. Consequently, both output signals become LOW, i.e., OUT1=0 and OUT2=0, and IPeak=IPeak1.
When the load current increases from below ILoad3 to above ILoad1, the operation of the circuit in FIG. 23 is similar to the operation described above just in the opposite direction.
It should be noted that references voltages VRef(ILoad1), VRef(ILoad2), and VRef(ILoad3), which correspond to load currents ILoad1, ILoad2, and ILoad3, respectively, are constant when the input voltage is constant. When the input voltage is variable, these references voltages vary inversely proportionally to the input voltage.
FIG. 25 presents one implementation method according to the embodiment presented in FIG. 12. The implementation method is obtained from the method presented in FIG. 10 by replacing the switching frequency with the feedback voltage because the feedback voltage is proportional to the switching frequency. The implementation according to the method presented in FIG. 25 is similar to the implementation presented in FIG. 13.
It will be understood that the above description of the present invention is susceptible to various modifications, changes and adaptations, and that the same are intended to be comprehended within the meaning and range of equivalents of the appended claims.

## Claims

1. A power supply that provides output voltage to a load comprising:
a power converter coupled to the load, said power converter having at least one switch that is operated at a variable switching frequency that decreases towards an audible frequency range as the load decreases, wherein current pulses are produced through the at least one switch in accordance with the variable switching frequency, said variable switching frequency initially being inside the audible frequency range when the load decreases below a first level;
a controller that regulates the output voltage by controlling said variable switching frequency based on a feedback signal coupled to the load, said controller comprising an error amplifier, a voltage-controlled oscillator, a comparator, a current sense resistor, and a switch timing and drive circuit, said comparator comparing a current sense voltage developed across the current sense resistor by the switch current pulses with a current-sense reference voltage; and
an audible noise reduction circuit that modifies said variable switching frequency of said power converter when said variable switching frequency is inside the audible frequency range, the audible noise reduction circuit configured to operate by:
increasing the variable switching frequency above a top threshold of he audible frequency range when the load is between the first level and a second level smaller than the first level,
increasing the variable switching frequency to an upper part of the audible frequency range that is below the top threshold of the audible frequency range when the load is between the second level and a third level smaller than the second level, and
decreasing the variable switching frequency to a lower part of the audible frequency range that is above a bottom threshold of the audible frequency range when the toad is below the third level.

2. The power supply of claim 1, wherein the audible noise reduction circuit, when the load is between said first level and said second level, in order to increase said variable switching frequency above the top threshold of the audible frequency range, reduces the peak value of the current pulses in one or more discrete steps so that in response to a decreasing load, every time the switching frequency decreases to the top threshold of the audible frequency range, the peak value of the current pulses is reduced by one step, resulting in an increase of said variable switching frequency in a discrete step above the top threshold of the audible frequency range.

3. The power supply of claim 1, wherein the audible noise reduction circuit, when the load is between said second level and said third level, in order to increase said variable switching frequency to be inside said upper part of the audible frequency range that is below the top threshold of the audible frequency range, reduces the peak value of the current pulses to a minimum level, so that the resulting audible noise is lower than a maximum acceptable level.

4. The power supply of claim 1, wherein the audible noise reduction circuit, when the load is below said third level, in order to decrease the variable switching frequency to a lower part of the audible frequency range that is above the bottom threshold of the audible frequency range, increases the peak value of the current pulses.

5. The power supply of claim 2, wherein the audible noise reduction circuit changes the peak value of current pulses based on monitoring a switch drive signal for the at least one switch, or monitoring a load current, or monitoring a feedback voltage.

6. The power supply of claim 3, wherein the audible noise reduction circuit changes the peak value of current pulses based on monitoring a switch drive signal for the at least one switch, or monitoring a load current, or monitoring a feedback voltage.

7. The power supply of claim 4, wherein the audible noise reduction circuit changes the peak value of current pulses based on monitoring a switch drive signal for the at least one switch, or monitoring a load current, or monitoring a feedback voltage.

8. The power supply of claim 5, wherein the peak value of the current pulses is changed by adjusting at least one of current-sense reference voltage, current-sense voltage, or current-sense resistor.

9. The power supply of claim 6, wherein the peak value of the current pulses is changed by adjusting at least one of current-sense reference voltage, current-sense voltage, or current-sense resistor.

10. The power supply of claim 7, wherein the peak value of the current pulses is changed by adjusting at least one of current-sense reference voltage, current-sense voltage, or current-sense resistor.

11. The power supply of claim 1, wherein said power converter is at least one of:
a forward converter;
a half-bridge converter;
a full-bridge converter;
a flyback converter; or
a boost converter.