# Patent

## Title
High speed latch comparators

## Abstract

In a latch circuit having a bistable pair of cross connected transistors of a first polarity and a third transistor of a second polarity, a current signal greater than a bias current is received at a latch circuit port, amplified with the third transistor, and applied to the latch circuit port. This decreases the time in which the latch circuit port receiving the current signal greater than the bias current reaches a steady state voltage.

## Background
CROSS-REFERENCE TO RELATED APPLICATIONS
This application is a continuation of U.S. application Ser. No. 12/040,805, filed Feb. 29, 2008, now U.S. Pat. No. 7,906,992, which is a divisional of U.S. application Ser. No. 10/649,808, filed Aug. 28, 2003, now U.S. Pat. No. 7,352,215, which is a continuation of U.S. application Ser. No. 10/083,463, filed Feb. 27, 2002, now U.S. Pat. No. 6,639,430, which claims the benefit of U.S. Provisional Application No. 60/271,425, filed Feb. 27, 2001, all of which are incorporated herein in its entirety by reference.


BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to high speed latch comparators.
2. Background Art
Commercialization of the Internet has proven to be a mainspring for incentives to improve network technologies. Development programs have pursued various approaches including strategies to leverage use of the existing Public Switched Telephone Network and plans to expand use of wireless technologies for networking applications. Both of these approaches (and others) entail the conversion of data between analog and digital formats. Therefore, it is expected that analog-to-digital converters (ADCs) and digital-to-analog converters (DACs) will continue to perform critical functions in many network applications.
Because ADCs find uses in a wide variety of applications, design of these circuits has evolved along many paths to yield several distinct architectures, including “delta sigma,” “successive approximation,” “pipelined,” and “flash.” Each architecture has its benefits and drawbacks. Paramount among these is a tradeoff between bandwidth and degree of resolution. FIG. 1 is a graph 100 that shows the tradeoff between bandwidth and degree of resolution for the various ADC architectures. Graph 100 comprises a “degree of resolution” axis 102 and a “bandwidth” axis 104. The relative positions of the different ADC architectures are plotted with respect to axes 102, 104: a “delta sigma” region 106, a “successive approximation” region 108, a “pipelined” region 110, and a “flash” region 112. In the design of network technologies, data conversion has often presented itself as a bottleneck that impedes the rate at which information is transmitted. Therefore, those ADC architectures that can support large bandwidths for rapid transfers of data have been favored for network applications.
FIG. 2A is a block diagram of an exemplary conventional two-bit flash ADC 200. ADC 200 comprises a first comparator “A” 202, a second comparator “B” 204, a third comparator “C” 206, a priority encoder 208, a first resistor “R1” 210, a second resistor “R2” 212, a third resistor “R3” 214, and a fourth resistor “R4” 216. Each of R1 210, R2 212, R3 214, and R4 216 has the same measure of resistance. R1 210, R2 212, R3 214, and R4 216 are connected in series between an analog ground “VAG” 218 and a supply voltage “V” 220. R1 210 is connected between VAG 218 and a first node “N1” 222. R2 212 is connected between N1 222 and a second node “N2” 224. R3 214 is connected between N2 224 and a third node “N3” 226. R4 216 is connected between N3 226 and V 220. In this configuration, the voltage at N1 222 is equal to V/4, the voltage at N2 224 is equal to V/2, and the voltage at N3 226 is equal to 3V/4.
The inverting terminals of comparators A 202, B 204, and C 206 are connected to, respectively, N1 222, N2 224, and N3 226. An analog signal “x” 228 is received at an input 230, which is connected to the noninverting terminals of comparators A 202, B 204, and C 206. A quantized signal is produced at the output terminal of each comparator. Quantized signals “w1” 232, “w2” 234, and “w3” 236 are produced at the output terminals of, respectively, comparators A 202, B 204, and C 206. Each quantized signal has a voltage with a value “LOW” or a value “HIGH” depending upon whether a corresponding value of the voltage of analog signal x 228 is less than (or equal to) or greater than the voltage at the inverting terminal of the corresponding comparator (i.e., the reference voltage of the comparator). For example, when the value of the voltage of analog signal x 228 is less than or equal to V/4, the values of the voltages of w3 236, w2 234, and w1 232 are equal to, respectively, LOW, LOW, and LOW. When the value of the voltage of analog signal x 228 is less than or equal to V/2, but greater than V/4, the values of the voltages of w3 236, w2 234, and w1 232 are equal to, respectively, LOW, LOW, and HIGH. When the value of the voltage of analog signal x 228 is less than or equal to 3V/4, but greater than V/2, the values of the voltages of w3 236, w2 234, and w1 232 are equal to, respectively, LOW, HIGH, and HIGH. When the value of the voltage of analog signal x 228 is less than or equal to V, but greater than 3V/4, the values of the voltages of w3 236, w2 234, and w1 232 are equal to, respectively, HIGH, HIGH, and HIGH. It is because quantized signals w1 232, w2 234, and w3 236 are produced simultaneously that two-bit flash ADC 200, also referred to as a “parallel-comparator” ADC, is capable of supporting large bandwidths for rapid transfers of data.
The output terminals of comparators A 202, B 204, and C 206 are connected to priority encoder 208. Quantized signals w1 232, w2 234, and w3 236 are received by priority encoder 208, which processes them to produce, at an output 238, a two-bit digital signal “y” comprising a least significant bit (LSB) signal “y1” 240 and a most significant bit (MSB) signal “y2” 242. FIG. 2B is a truth table 244 for priority encoder 208. In truth table 244, LOW and HIGH are encoded as, respectively, 0 and 1. When quantized signals w3 236, w2 234, and w1 232 are equal to, respectively, 0, 0, and 0, bit signals y2 242 and y1 240 are equal to, respectively, 0 and 0, which corresponds to binary number 0. When quantized signals w3 236, w2 234, and w1 232 are equal to, respectively, 0, 0, and 1, bit signals y2 242 and y1 240 are equal to, respectively, 0 and 1, which corresponds to binary number 1. When quantized signals w3 236, w2 234, and w1 232 are equal to, respectively, 0, 1, and 1, bit signals y2 242 and y1 240 are equal to, respectively, 1 and 0, which corresponds to binary number 2. When quantized signals w3 236, w2 234, and w1 232 are equal to, respectively, 1, 1, and 1, bit signals y2 242 and y1 240 are equal to, respectively, 1 and 1, which corresponds to binary number 3.
The skilled artisan will appreciate that, with additional comparators and resistors and by using a priority encoder capable of processing additional quantized signals, flash ADC 200 can be modified so that digital signal y comprises more than two bit signals. Alternatively, flash ADC 200 can be modified so that digital signal y comprises one bit signal.
Implementations of comparators A 202, B 204, and C 206 often use latch circuits, and are referred to as latch comparators. FIG. 3 is a schematic diagram of an exemplary conventional latch circuit 300 that can be used in an implementation of any of comparators A 202, B 204, or C 206. Latch circuit 300 comprises a bistable pair 302 connected between a reset switch 304 and analog ground VAG 218. (Alternatively, analog ground VAG 218 can be replaced by a first supply voltage “VSS”.) Preferably, bistable pair 302 comprises a first NMOSFET (n-channel Metal Oxide Semiconductor Field Effect Transistor) “M1” 306 and a second NMOSFET “M2” 308. Ideally, M1 306 and M2 308 are matched transistors. Preferably, each of M1 306 and M2 308 has a gain greater than one. However, bistable pair 302 can function if the product of the individual gains of M1 306 and M2 308 (i.e., the loop gain) is greater than one. The gate terminal of M2 308 is connected to the drain terminal of M1 306 at a first port “N4” 310. The gate terminal of M1 306 is connected to the drain terminal of M2 308 at a second port “N5” 312. The source terminals of M1 306 and M2 308 are together connected to analog ground VAG 218. In this configuration, M1 306 and M2 308 are said to be cross connected. Preferably, reset switch 304 comprises a third NMOSFET “M3” 314. The source terminal of M3 314 is connected to the drain terminal of one of M1 306 or M2 308; the drain terminal of M3 314 is connected the drain terminal of the other of M1 306 or M2 308. A clock waveform “Ck” 316 is applied to the gate terminal of M3 314. Ck 316 cycles between an “UP” voltage and an “DOWN” voltage at a sampling frequency.
The skilled artisan will appreciate that M1 306, M2 308, and M3 314 can also be realized in other field effect, junction, or combination transistor technologies. In general, reset switch 304 can be realized in a variety of switch technologies, including microelectromechanical embodiments. Latch circuit 300 can also be used for other applications.
For each latch circuit 300 in ADC 200, quantized signal “w” (e.g., w1 232, w2 234, or w3 236) is produced as an output voltage at N4 310 or N5 312. Latch circuit 300 is often preceded by an input stage (not shown) that includes a differential amplifier so that the voltage of analog signal x 228, applied at the noninverting terminal of the comparator, can be compared with the voltage at the inverting terminal of the comparator. For example, the voltage of analog signal x 228 is compared with V/4, for comparator A 202; V/2, for comparator B 204; and 3V/4, for comparator C 206.
The input stage produces a differential current signal comprising a first current signal “i1” 318 and a second current signal “i2” 320. First and second current signals i1 318 and i2 320 each comprise a bias current “ib” and a signal current “is”. The relationship between bias current ib and signal current is in first current signal i1 318 can be expressed as shown in Eq. (1):

i1=ib+(1⁄2)(is),  Eq. (1)

while the relationship between bias current ib and signal current is in second current signal i2 320 can be expressed as shown in Eq. (2):

i2=ib−(1⁄2)(is).  Eq. (2)

The differential amplifier is configured so that first current signal i1 318 increases and decreases in response to, respectively, the rise and drop of the voltage of analog signal x 228, while second current signal i2 320 increases and decreases in response to, respectively, the drop and rise of the voltage of analog signal x 228. Thus, first and second current signals i1 318 and i2 320 always change currents in opposite directions, but the sum of first and second current signals i1 318 and i2 320 remains constant. In latch circuit 300, first current signal i1 318 and second current signal i2 320 are received as input current signals at, respectively, N4 310 and N5 312.
In latch circuit 300, when the voltage of Ck 316 is UP (i.e, the reset phase), M3 314 connects N4 310 with N5 312, so that the steady state voltages at both nodes are equal, and bias current ib flows through each of M1 306 and M2 308. Parasitic capacitances at each of nodes N4 310 and N5 312 are charged by bias current ib that flows through each of M1 306 and M2 308. The skilled artisan will appreciate that the parasitic capacitance at, for example, N4 310, includes the gate-to-source capacitance of M2 308, the drain-to-substrate capacitance of M1 306, the drain-to-substrate capacitance of M3 314, and the capacitance of the wiring connecting circuit devices. Bias current ib charges the parasitic capacitances at each of nodes N4 310 and N5 312 so that the voltages at N4 310 and N5 312 are at a “MID” value that is between LOW and HIGH. The gate and drain terminals of M1 306 and M2 308 are connected together. M1 306 and M2 308 are sized so that, under these conditions, they operate in “ON” states.
When the voltage of Ck 316 is DOWN (i.e., the sampling phase), the states of M1 306 and M2 308 are controlled by first and second current signals i1 318 and i2 320. For example, when first current signal i1 318 is greater than bias current ib and second current signal i2 320 is less than bias current ib, a transient is initiated to force M1 306 to operate in an “OFF” state, while M2 308 remains operating in an ON state. The course of this transient depends on how first and second current signals i1 318 and i2 320 change during the sampling phase. However, if M1 306 is turned OFF and the parasitic capacitances at N4 310 are fully charged by first current signal i1 318 (i.e., at a new steady state), the voltage at N4 310 is HIGH and the voltage at N5 312 is LOW. The transient can be explained in two parts. The first part describes the changes that occur while M1 306 remains ON. The second part depicts the conclusion of the excursion after M1 306 is turned OFF.
When first current signal i1 318 is greater than bias current ib, first current signal i1 318 continues to charge the parasitic capacitances at N4 310, which causes the voltage at N4 310 to rise. This is indicated by a small up-arrow “a” 322. Contemporaneously, when second current signal i2 320 is less than bias current ib, the parasitic capacitances at N5 312 start to discharge, which causes the voltage at N5 312 to drop. This is indicated by a small down-arrow “b” 324.
Because the voltage at N4 310 is also the voltage at the gate terminal of M2 308, the voltage at the gate terminal of M2 308 rises by the same amount as the rise in the voltage at N4 310. This is indicated by a small up-arrow “c” 326, where small up-arrow c 326 has the same length (i.e., the same change in voltage) as small up-arrow a 322. Because the voltage at the source terminal of M2 308 is held at analog ground VAG 218, the gate-to-source voltage of M2 308 increases by the same amount as the rise in the voltage at the gate terminal of M2 308. The increase in the gate-to-source voltage of M2 308 causes its drain current to increase. In response to the increase in the gate-to-source voltage of M2 308 and the increase in its drain current, the drain-to-source voltage of M2 308 decreases by a greater magnitude than the increase in its gate-to-source voltage. This is indicated by a large down-arrow “d” 328, where large down-arrow d 328 has a longer length (i.e., a larger change in voltage) than small up-arrow c 326. Because the voltage at the source terminal of M2 308 is held at analog ground VAG 218, the voltage at N5 312 drops by the same amount as the decrease in drain-to-source voltage of M2 308. Thus, the voltage at N5 312 drops under the relatively small effect of second current signal i2 320 being less than bias current ib (i.e., small down-arrow b324), and the relatively large effect of the decrease in the drain-to-source voltage of M2 308 (i.e., large down-arrow d 328).
Likewise, because the voltage at N5 312 is also the voltage at the gate terminal of M1 306, the voltage at the gate terminal of M1 306 drops by the same amount as the drop in the voltage at N5 312. This is indicated by a small down-arrow “e” 330, where small down-arrow e 330 has the same length (i.e., the same change in voltage) as small down-arrow b 324. Because the voltage at the source terminal of M1 306 is held at analog ground VAG 218, the gate-to-source voltage of M1 306 decreases by the same amount as the drop in the voltage at the gate terminal of M1 306. The decrease in the gate-to-source voltage of M1 306 causes its drain current to decrease. In response to the decrease in the gate-to-source voltage of M1 306 and the decrease in its drain current, the drain-to-source voltage of M1 306 increases by a greater magnitude than the decrease in its gate-to-source voltage. This is indicated by a large up-arrow “f” 332, where large up-arrow f 332 has a longer length (i.e., a larger change in voltage) than small down-arrow e 330. Because the voltage at the source terminal of M1 306 is held at analog ground VAG 218, the voltage at N4 310 rises by the same amount as the increase in drain-to-source voltage of M1 306. Thus, the voltage at N4 310 rises under the relatively small effect of first current signal i1 318 being greater than bias current ib (i.e., small up-arrow a 322) and the relatively large effect of the increase in the drain-to-source voltage of M1 306 (i.e., large up-arrow f 332).
The increasing of the drain-to-source voltage of M1 306 and the decreasing of the drain-to-source voltage of M2 308 reinforce each other. The gate-to-source voltage of M1 306 decreases with the drain-to-source voltage of M2 308 until M1 306 is turned OFF.
When M1 306 is OFF, it does not conduct current. Without drain current, the decreasing of the gate-to-source voltage of M1 306 no longer effects its drain-to-source voltage. Thus, the voltage at N4 310 continues to rise solely under the relatively small effect of first current signal i1 318 being greater than bias current ib (i.e., small up-arrow a 322) until the parasitic capacitances at N4 310 are fully charged and the voltage at N4 310 is HIGH.
However, because the voltage at N4 310 is also the voltage at the gate terminal of M2 308, the voltage at the gate terminal of M2 308 continues to rise. Because M2 308 remains ON, the increase in its gate-to-source voltage causes the drain current of M2 308 to increase, which in turn causes its drain-to-source voltage to decrease by a greater magnitude than the increase in the gate-to-source voltage of M2 308. Thus, the voltage at N5 312 continues to drop under the relatively small effect of second current signal i2 320 being less than bias current ib (i.e., small down-arrow b 324) and the relatively large effect of the decrease in the drain-to-source voltage of M2 308 (i.e., large down-arrow d 328) until the discharge of the parasitic capacitances at N5 312 is balanced and the voltage at N5 312 is LOW.
Therefore, it is a characteristic of latch circuit 300 that the port (i.e., N4 310 or N5 312) receiving the current signal (i.e., i1 318 or i2 320) that is greater than bias current ib requires more time to reach its new steady state voltage than the port receiving the current signal that is less than bias current ib. In practical implementations of latch circuit 300, the port receiving the current signal that is greater than bias current ib can require three to five times as much time to reach its new steady state voltage as that of the port receiving the current signal that is less than bias current ib. This limitation determines the frequency of Ck 316, and ultimately the processing speed of ADC 200.
Furthermore, if first and second current signals i1 318 and i2 320 both have values near to that of bias current ib (i.e., small signal current is), it is possible that the output voltage (at N4 310 or N5 312) may not reach LOW or HIGH before the end of the sampling phase. In this situation, ADC 200 does not produce a digital output. Such a “non-decision” is referred to as a “bit error”. Bit errors can adversely effect the performance of a system that uses the digital output of ADC 200. Such systems typically require bit error rates on an order of 10−18 to 10−16. Traditionally, bit errors are reduced by cascading latch comparators, where the overall bit error rate of the system is the product of the bit error rate of each cascaded latch comparator. However, this solution delays processing, complicates circuit design, uses additional die area, and consumes more power. Thus, there is a need to decrease the time necessary for the port (i.e., N4 310 or N5 312) receiving the current signal (i.e., i1 318 or i2 320) that is greater than bias current ib to reach its new steady state voltage.
BRIEF SUMMARY OF THE INVENTION
The present invention relates to high speed latch comparators. In a latch circuit having a bistable pair of cross connected transistors of a first polarity and a third transistor of a second polarity, a current signal greater than a bias current is received at a latch circuit port, amplified with the third transistor, and applied to the latch circuit port. This decreases the time in which the latch circuit port receiving the current signal greater than the bias current reaches a steady state voltage. Advantageously, the time in which the latch circuit port receiving the current signal less than the bias current reaches a steady state voltage also decreases.
In an embodiment, a latch circuit comprises a bistable pair and a vertical latch. The bistable pair is connected between a reset switch and a first supply voltage. The bistable pair has a first port capable of receiving a first current signal and producing a first output voltage, and a second port capable of receiving a second current signal and producing a second output voltage. The vertical latch is connected between the first supply voltage and a second supply voltage, and connected to the bistable pair. The vertical latch acts to decrease the time necessary for the port receiving the current signal that is greater than bias current ib to reach its new steady state voltage.
Preferably, the bistable pair comprises a first MOSFET and a second MOSFET such that the gate terminal of the first MOSFET is connected to the drain terminal of the second MOSFET, the gate terminal of the second MOSFET is connected to the drain terminal of said the MOSFET, and the source terminals of the first and the second MOSFETs are connected to the first supply voltage.
Preferably, the vertical latch comprises a first MOSFET current mirror pair connected to the bistable pair, and a second MOSFET current mirror pair connected to the first MOSFET current mirror pair. The first current mirror can comprise a third MOSFET connected to the first MOSFET of the bistable pair. The second current mirror can comprise a fourth MOSFET connected to the third MOSFET. The source terminal of the third MOSFET is connected to the first supply voltage. The drain terminal of the third MOSFET is connected to the gate terminal of the fourth MOSFET. The source terminal of the fourth MOSFET is connected to the second supply voltage. The drain terminal of the fourth MOSFET is connected to the gate terminal of the third MOSFET, the gate terminal of the first MOSFET, and the drain terminal of the second MOSFET.
The first current mirror can further comprise a fifth MOSFET connected to the third MOSFET. The second current minor can further comprise a sixth MOSFET connected to the fourth MOSFET. The source terminal of the fifth MOSFET is connected to the first supply voltage. The drain terminal of the fifth MOSFET is connected to the drain terminal of the fourth MOSFET. The gate terminal of the fifth MOSFET is connected to the gate terminal of the third MOSFET and the drain terminal of the fifth MOSFET. The source terminal of the sixth MOSFET is connected to the second supply voltage. The drain terminal of the sixth MOSFET is connected to the drain terminal of the third MOSFET. The gate terminal of the sixth MOSFET is connected to the gate terminal of the fourth MOSFET and the drain terminal of the sixth MOSFET.
The present invention further comprises a method for reducing the power consumed by the latch circuit. When the bistable pair and the vertical latch are reset, the third or fourth MOSFET can be held OFF. After resetting, the fourth MOSFET can be held OFF when the second MOSFET changes state from ON to OFF.
In an embodiment, the fourth MOSFET can be held OFF during the reset phase by a vertical latch reset switch connected to the vertical latch. The vertical latch reset switch can comprise a MOSFET connected between the second supply voltage and the vertical latch. During the reset phase, the vertical latch reset switch connects the gate terminal of the fourth MOSFET to the second supply voltage. Preferably, a voltage source is connected between the vertical latch reset switch and the second supply voltage to decrease the time in which the vertical latch reset switch turns OFF, which decreases the time in which the vertical latch can act to decrease the time necessary for the port receiving the current signal that is greater than bias current ib to reach its new steady state voltage.
In a related embodiment, the latch circuit can further comprise a second vertical latch connected between the first supply voltage and the second supply voltage, and connected to the bistable pair at the second MOSFET of the bistable pair. A second vertical latch reset switch can be connected to the second vertical latch. A second voltage source can be connected between the second vertical latch reset switch and the second supply voltage.
In a further related embodiment, the fourth MOSFET can be held OFF after resetting when the second MOSFET changes state from ON to OFF by a second bistable pair connected to the second supply voltage, the vertical latch, and the second vertical latch. Preferably, the second bistable pair comprises a first MOSFET and a second MOSFET such that the gate terminal of the first MOSFET is connected to the drain terminal of the second MOSFET, the gate terminal of the second MOSFET is connected to the drain terminal of said the MOSFET, and the source terminals of the first and the second MOSFETs are connected to the second supply voltage. The drain terminal of the second MOSFET of the second bistable pair is connected to the gate terminal of the fourth MOSFET of the vertical latch. After resetting, when the second MOSFET changes state from ON to OFF, the second MOSFET of the second bistable pair turns ON and holds the fourth MOSFET OFF.
In yet a further related embodiment, both the third and fourth MOSFETs can be held OFF during the reset phase by replacing the vertical latch reset switch and the second vertical latch reset switch with a third vertical latch reset switch connected to the second bistable pair, the vertical latch, and the second vertical latch. The third vertical latch reset switch can comprise a MOSFET connected between the second bistable pair and the second supply voltage, connected between the vertical latch and the second supply voltage, and connected between the second vertical latch and the second supply voltage. During the reset phase, the third vertical latch reset switch disconnects the fourth MOSFET from the second supply voltage without connecting the third MOSFET to the second supply voltage.
In an alternative embodiment, the third MOSFET can be held OFF during the reset phase by a reset circuit connected to the bistable pair, the vertical latch, and the second vertical latch. The reset circuit can comprise a first MOSFET connected between the bistable pair and the first supply voltage, and a second MOSFET connected between the vertical latch and the first supply voltage. During the reset phase, the second MOSFET of the reset circuit disconnects the third MOSFET from the first supply voltage.
The reset switch can comprise a MOSFET connected between the first port and the second port. A clock voltage is applied to the gate terminal of the MOSFET. The present invention further comprises a method for reducing the clock voltage where the reset switch comprises a reset circuit.
In an embodiment, the reset circuit comprises a first MOSFET, a second MOSFET, and a third MOSFET. The first MOSFET is connected to the first supply voltage. The second MOSFET is connected between the first MOSFET and the first port. The third MOSFET is connected between the first MOSFET and the second port. The clock voltage is applied to the gate terminal of the first MOSFET.
The present invention also includes a comparator comprising an input stage, a latch circuit, and an output stage. The input stage is capable of receiving an analog signal. The latch circuit is connected to the input stage. The latch circuit has a bistable pair and a vertical latch. The output stage is connected to the latch circuit. The output stage is capable of retaining an output of the latch circuit.



BRIEF DESCRIPTION OF THE FIGURES
The accompanying drawings, which are incorporated herein and form part of the specification, illustrate the present invention and, together with the description, further serve to explain the principles of the invention and to enable a person skilled in the pertinent art to make and use the invention.
FIG. 1 is a graph 100 that shows the tradeoff between bandwidth and degree of resolution for the various ADC architectures.
FIG. 2A is a block diagram of an exemplary conventional two-bit flash ADC 200.
FIG. 2B is a truth table 244 for priority encoder 208.
FIG. 3 is a schematic diagram of an exemplary conventional latch circuit 300 that can be used in an implementation of any of comparators A 202, B 204, or C 206.
FIG. 4 is a schematic diagram of a latch circuit 400 that explains the present invention.
FIG. 5 is a schematic diagram of a latch circuit 500, in which latch circuit 400 further comprises a vertical latch reset switch 502.
FIG. 6 is a schematic diagram of a latch circuit 600, in which latch circuit 500 further comprises a second vertical latch 602 and a second vertical latch reset switch 604.
FIG. 7 is a schematic diagram of a latch circuit 700, in which latch circuit 600 further comprises a second bistable pair 702.
FIG. 8 is a schematic diagram of a latch circuit 800, in which reset switch 304 of latch circuit 700 is replaced by a reset circuit 802.
FIG. 9 shows a flow chart of a method 900 for decreasing the time in which a latch circuit port receiving a current signal greater than a bias current reaches a steady state voltage.
FIG. 10 is a schematic diagram of a latch circuit 1000, in which latch circuit 700 further comprises a reset circuit 1002.
FIG. 11 is a schematic diagram of a latch circuit 1100, in which latch circuit 400 further comprises reset circuit 1002.
FIG. 12 is a schematic diagram of a latch circuit 1200, in which latch circuit 500 further comprises a voltage source “ΔV1” 1202.
FIG. 13 is a schematic diagram of a latch circuit 1300, in which latch circuit 700 further comprises voltage source ΔV1 1202 and a second voltage source “ΔV2” 1302.
FIG. 14 is a schematic diagram of a latch circuit 1400, in which vertical latch reset switch 502 and second vertical latch reset switch 604 of latch circuit 700 are replaced by a third vertical latch reset switch 1402.
FIG. 15 is a graph 1500 of a current 1502 drawn by latch circuit 700 as a function of time “t” 1504 after the voltage of Ck 316 changes from UP to DOWN.
FIG. 16 is a schematic diagram of a comparator 1600 implemented using latch circuit 700.
FIG. 17 is a schematic diagram of an alternative embodiment 1700 of vertical latch 402.
FIG. 18 shows a flow chart of a method 1800 for reducing the power consumed by a latch circuit.



The preferred embodiments of the invention are described with reference to the figures where like reference numbers indicate identical or functionally similar elements. Also in the figures, the left-most digit(s) of each reference number identify the figure in which the reference number is first used.
DETAILED DESCRIPTION OF THE INVENTION
The present invention relates to high speed latch comparators. FIG. 4 is a schematic diagram of a latch circuit 400 that explains the present invention. Latch circuit 400 comprises bistable pair 302 connected between reset switch 304 and analog ground VAG 218. (Alternatively, analog ground VAG 218 can be replaced by first supply voltage VSS.) Preferably, bistable pair 302 comprises first NMOSFET M1 306 and second NMOSFET M2 308 connected in the same manner as in latch circuit 300. Ideally, M1 306 and M2 308 are matched transistors. Preferably, each of M1 306 and M2 308 has a gain greater than one. However, bistable pair 302 can function if the product of the individual gains of M1 306 and M2 308 (i.e., the loop gain) is greater than one. Preferably, reset switch 304 comprises third NMOSFET M3 314. Clock waveform Ck 316 is applied to the gate terminal of M3 314.
A vertical latch 402 is connected between analog ground VAG 218 and a second supply voltage “VDD” 404. Preferably, vertical latch 402 comprises a fourth NMOSFET “M4” 406 and a first PMOSFET (p-channel MOSFET) “M5” 408. Ideally, M4 406 and M5 408 are matched transistors. Preferably, each of M4 406 and M5 408 has a gain greater than one. However, vertical latch 402 can function if the product of the individual gains of M4 406 and M5 408 (i.e., the loop gain) is greater than one. The source terminal of M4 406 is connected to analog ground VAG 218. The drain terminal of M4 406 is connected to the gate terminal of M5 408. The gate terminal of M4 406 is connected to the gate terminal of M2 308. The source terminal of M5 408 is connected to second supply voltage VDD 404. The drain terminal of M5 408 is connected to the gate terminal of M4 406. The skilled artisan will appreciate that M4 406 and M5 408 can also be realized in other field effect, junction, or combination transistor technologies.
In latch circuit 400, when the voltage of Ck 316 is LIP (i.e, the reset phase), M3 314 connects N4 310 with N5 312, so that the steady state voltages at both nodes are equal, and bias current ib flows through each of M1 306 and M2 308. Parasitic capacitances at each of nodes N4 310 and N5 312 are charged by bias current ib that flows through each of M1 306 and M2 308. Bias current ib charges the parasitic capacitances at each of nodes N4 310 and N5 312 so that the voltages at N4 310 and N5 312 are at MID. The gate and drain terminals of M1 306 and M2 308 are connected together. M1 306 and M2 308 are sized so that, under these conditions, they are both ON.
When the voltage of Ck 316 is DOWN (i.e., the sampling phase), the states of Mi 306 and M2 308 are controlled by first and second current signals 318 and i2 320. For example, when first current signal i1 318 is greater than bias current ib and second current signal i2 320 is less than bias current ib, a transient is initiated to force M1 306 OFF, while M2 308 remains ON. The course of this transient depends on how first and second current signals i1 318 and i2 320 change during the sampling phase. However, if M1 306 is turned OFF and the parasitic capacitances at N4 310 are fully charged by first current signal i1 318 (i.e., at a new steady state), the voltage at N4 310 is HIGH and the voltage at N5 312 is LOW. The transient can be explained in two parts. The first part describes the changes that occur while M1 306 remains ON. The second part depicts the conclusion of the excursion after M1 306 is turned OFF.
When first current signal i1 318 is greater than bias current ib, first current signal i1 318 continues to charge the parasitic capacitances at N4 310, which causes the voltage at N4 310 to rise. This is indicated by small up-arrow a 322. Contemporaneously, when second current signal i2 320 is less than bias current ib, the parasitic capacitances at N5 312 start to discharge, which causes the voltage at N5 312 to drop. This is indicated by small down-arrow b 324.
Because the voltage at N5 312 is also the voltage at the gate terminal of M1 306, the voltage at the gate terminal of M1 306 drops by the same amount as the drop in the voltage at N5 312. This is indicated by small down-arrow e 330, where small down-arrow e 330 has the same length (i.e., the same change in voltage) as small down-arrow b 324. Because the voltage at the source terminal of M1 306 is held at analog ground VAG 218, the gate-to-source voltage of M1 306 decreases by the same amount as the drop in the voltage at the gate terminal of M1 306. The decrease in the gate-to-source voltage of M1 306 causes its drain current to decrease. In response to the decrease in the gate-to-source voltage of M1 306 and the decrease in its drain current, the drain-to-source voltage of M1 306 increases by a greater magnitude than the decrease in its gate-to-source voltage. This is indicated by large up-arrow f 332, where large up-arrow f 332 has a longer length (i.e., a larger change in voltage) than small down-arrow e 330.
Meanwhile, because the voltage at N4 310 is also the voltage at the gate terminal of M2 308, the voltage at the gate terminal of M2 308 rises by the same amount as the rise in the voltage at N4 310. This is indicated by small up-arrow c 326, where small up-arrow c 326 has the same length (i.e., the same change in voltage) as small up-arrow a 322. Likewise, because the voltage at N4 310 is also the voltage at the gate terminal of M4 406, the voltage at the gate terminal of M4 406 rises by the same amount as the rise in the voltage at N4 310. This is indicated by a small up-arrow “g” 410, where small up-arrow g 410 has the same length (i.e., the same change in voltage) as small up-arrow a 322.
Because the voltage at the source terminal of M2 308 is held at analog ground VAG 218, the gate-to-source voltage of M2 308 increases by the same amount as the rise in the voltage at the gate terminal of M2 308. The increase in the gate-to-source voltage of M2 308 causes its drain current to increase. In response to the increase in the gate-to-source voltage of M2 308 and the increase in its drain current, the drain-to-source voltage of M2 308 decreases by a greater magnitude than the increase in its gate-to-source voltage. This is indicated by large down-arrow d 328, where large down-arrow d 328 has a longer length (i.e., a larger change in voltage) than small up-arrow c 326. Likewise, because the voltage at the source terminal of M4 406 is held at analog ground VAG 218, the gate-to-source voltage of M4 406 increases by the same amount as the rise in the voltage at the gate terminal of M4 406. The increase in the gate-to-source voltage of M4 406 causes its drain current to increase. In response to the increase in the gate-to-source voltage of M4 406 and the increase in its drain current, the drain-to-source voltage of M4 406 decreases by a greater magnitude than the increase in its gate-to-source voltage. This is indicated by a large down-arrow “h” 412, where large down-arrow h 412 has a longer length (i.e., a larger change in voltage) than small up-arrow c 326.
Because the voltage at the source terminal of M4 406 is held at analog ground VAG 218, the decrease in the drain-to-source voltage of M4 406 causes the voltage at the drain terminal of M4 406 to drop by the same amount. Because the voltage at the drain terminal of M4 406 is also the voltage at the gate terminal of M5 408, the voltage at the gate terminal of M5 408 drops by the same amount as the drop in the voltage at the drain terminal of M4 406. This is indicated by a large down-arrow “j” 414, where large down-arrow j 414 has the same length (i.e., the same change in voltage) as large down-arrow h 412. Because the voltage at the source terminal of M5 408 is held at second supply voltage VDD 404, the drop in the voltage at the gate terminal of M5 408 (i.e., a PMOSFET) causes its gate-to-source voltage to increase by the same amount. The increase in the gate-to-source voltage of M5 408 causes its drain current to increase. In response to the increase in the gate-to-source voltage of M5 408 and the increase in its drain current, the drain-to-source voltage of M5 408 decreases by a greater magnitude than the increase in its gate-to-source voltage. This is indicated by a larger up-arrow “k” 416, where larger up-arrow k 416 has a longer length (i.e., a larger change in voltage) than large down-arrow j 414.
Because the voltage at the source terminal of M2 308 is held at analog ground VAG 218, the voltage at N5 312 drops by the same amount as the decrease in drain-to-source voltage of M2 308. Thus, the voltage at N5 312 drops under the relatively small effect of second current signal i2 320 being less than bias current ib (i.e., small down-arrow b324), and the relatively large effect of the decrease in the drain-to-source voltage of M2 308 (i.e., large down-arrow d 328).
Because the voltage at N4 310 is also the voltage at the drain terminal of M5 408 and because the voltage at the source terminal of M5 408 is held at second supply voltage VDD 404, the voltage at N4 310 rises by the same amount as the decrease in the drain-to-source voltage of M5 408. Furthermore, because the voltage at the source terminal of M1 306 is held at analog ground VAG 218, the voltage at N4 310 rises by the same amount as the increase in drain-to-source voltage of M1 306. Additionally, because the voltage at the source terminal of M5 408 is held at second supply voltage VDD 404, the voltage at N4 310 also rises by the same amount as the decrease in drain-to-source voltage of M5 408 (i.e., a PMOSFET). Thus, the voltage at N4 310 rises under the relatively small effect of first current signal i1 318 being greater than bias current ib (i.e., small up-arrow a 322), the relatively large effect of the increase in the drain-to-source voltage of M1 306 (i.e., large up-arrow f 332), and the relatively larger effect of the decrease in the drain-to-source voltage of M5 408 (i.e., larger up-arrow k 416).
The increasing of the drain-to-source voltage of M1 306 and the decreasing of the drain-to-source voltage of M2 308 reinforce each other. The gate-to-source voltage of M1 306 decreases with the drain-to-source voltage of M2 308 until M1 306 is turned OFF.
When M1 306 is OFF, it does not conduct current. Without drain current, the decreasing of the gate-to-source voltage of M1 306 no longer effects its drain-to-source voltage. However, the voltage at N4 310 continues to rise under the relatively small effect of first current signal i1 318 being greater than bias current ib (i.e., small up-arrow a 322) and the relatively larger effect of the decrease in the drain-to-source voltage of M5 408 (i.e., larger up-arrow k 416) until the parasitic capacitances at N4 310 are fully charged and the voltage at N4 310 is HIGH.
It will be recognized that M4 406 and M5 408 foam a positive feedback loop that amplifies first current signal i1 318 and applies an exponentially growing current to the drain terminal of M1 306. Thus, the parasitic capacitances at N4 310 are charged under the combined effects of first current signal i1 318 and the exponentially growing current drawn from second supply voltage VDD 404 by M5 408.
Contemporaneously, because the voltage at N4 310 is also the voltage at the gate terminal of M2 308, the voltage at the gate terminal of M2 308 continues to rise. Because M2 308 remains ON, the increase in its gate-to-source voltage causes the drain current of M2 308 to increase, which in turn causes its drain-to-source voltage to decrease by a greater magnitude than the increase in the gate-to-source voltage of M2 308. Thus, the voltage at N5 312 continues to drop under the relatively small effect of second current signal i2 320 being less than bias current ib (i.e., small down-arrow b 324) and the relatively large effect of the decrease in the drain-to-source voltage of M2 308 (i.e., large up-arrow f 332) until the discharge of the parasitic capacitances at N5 312 is balanced and the voltage at N5 312 is LOW.
Thus, vertical latch 402 acts (i.e., larger up-arrow k 416) to decrease the time necessary for the port (i.e., N4 310) receiving the current signal (i.e., i1 318) that is greater than bias current ib to reach its new steady state voltage. Advantageously, the time necessary for the port (i.e., N4 310) receiving the current signal (i.e., i1 318) that is less than bias current ib to reach a steady state voltage also is decreased.
FIG. 5 is a schematic diagram of a latch circuit 500, in which latch circuit 400 further comprises a vertical latch reset switch 502. Preferably, vertical latch reset switch 502 comprises a second PMOSFET “M6” 504. The source terminal of M6 504 is connected to second supply voltage VDD 404. The drain terminal of M6 504 is connected to the gate terminal of M5 408. An inverse clock waveform “Ck.bar” 506 is applied to the gate terminal of M6 504. Ck.bar 506 cycles between DOWN voltage and UP voltage at the sampling frequency in a manner such that when the voltage of Ck 316 is UP, the voltage of Ck.bar 506 is DOWN, and vice versa. The skilled artisan will appreciate that M6 504 can also be realized in other field effect, junction, or combination transistor technologies. In general, vertical latch reset switch 502 can be realized in a variety of switch technologies, including microelectromechanical embodiments.
When the voltage of Ck.bar 506 is DOWN (i.e., the reset phase), M6 504 connects the gate terminal of M5 408 to second supply voltage VDD 404. With the gate and source terminals of M5 408 connected together, the gate-to-source voltage of M5 408 is made to equal zero, holding M5 408 OFF. This disrupts the latching action of vertical latch 402 so that bistable pair 302 can assume a state independent of the state of vertical latch 402.
When the voltage of Ck.bar 506 is DOWN, the voltage of Ck 316 is UP. When the voltage of Ck 316 is UP, M3 314 connects N4 310 and N5 312, so that the steady state voltages at both nodes are equal. Latch circuit 500 is configured so that the voltage at N4 310 equals the voltage at the drain terminal of M5 408. Because vertical latch reset switch 502 holds M5 408 OFF, it does not act to effect the state of bistable pair 302. Thus, with the gate and drain terminals of M1 306 and M2 308 connected together, latch circuit 500 is configured so that both M1 306 and M2 308 are ON. Because the gate-to-source voltage of M2 308 is also the gate-to-source voltage of M4 406, and M6 504 connects the drain terminal of M4 406 to second supply voltage VDD 404 such that the drain-to-source voltage of M6 504 is larger than that of M2 308, M4 406 is also ON.
Thus, by holding M5 408 OFF, vertical latch reset switch 502 reduces the power consumed by vertical latch 402 during the reset phase.
FIG. 6 is a schematic diagram of a latch circuit 600, in which latch circuit 500 further comprises a second vertical latch 602 and a second vertical latch reset switch 604. Second vertical latch 602 is connected between analog ground VAG 218 and second supply voltage VDD 404. Preferably, second vertical latch 602 comprises a fifth NMOSFET “M7” 606 and a third PMOSFET “M8” 608. Ideally, M7 606 and M8 608 are matched transistors. Preferably, each of M7 606 and M8 608 has a gain greater than one. However, second vertical latch 602 can function if the product of the individual gains of M7 606 and M8 608 (i.e., the loop gain) is greater than one. The source terminal of M7 606 is connected to analog ground VAG 218. The drain terminal of M7 606 is connected to the gate terminal of M8 608. The gate terminal of M7 606 is connected to the gate terminal of M1 306. The source terminal of M8 608 is connected to second supply voltage VDD 404. The drain terminal of M8 608 is connected to the gate terminal of M7 606. Preferably, second vertical latch reset switch 604 comprises a fourth PMOSFET “M9” 610. The source terminal of M9 610 is connected to second supply voltage VDD 404. The drain terminal of M9 610 is connected to the gate terminal of M8 608. Inverse clock waveform Ck.bar 506 is applied to the gate terminal of M9 610. The skilled artisan will appreciate that M7 606, M8 608, and M9 610 can also be realized in other field effect, junction, or combination transistor technologies. In general, second vertical latch reset switch 604 can be realized in a variety of switch technologies, including microelectromechanical embodiments.
Second vertical latch 602 and second vertical latch reset switch 604 operate in the same manner as vertical latch 402 and vertical latch reset switch 502. Whereas vertical latch reset switch 502 disrupts the latching action of vertical latch 402, second vertical latch reset switch 604 disrupts the latching action of second vertical latch 602. Likewise, while vertical latch 402 acts to decrease the time necessary for the port (i.e., N4 310) receiving the current signal (i.e., i1 318) that is greater than bias current ib to reach its new steady state voltage, second vertical latch 602 also acts to decrease the time necessary for the port (i.e., N5 312) receiving the current signal (i.e., i2 320) that is greater than bias current ib to reach its new steady state voltage. Because the time in which either port (i.e., N4 310 or N5 312) receiving the current signal (i.e., i1 318 or i2 320) that is greater than bias current ib reaches its new steady state voltage is decreased, latch circuit 600 is faster than latch circuit 300. This enables the frequency of Ck 316 (and Ck.bar 506) to be increased, which increases the processing speed of an ADC that incorporates latch circuit 600. Alternatively, where Ck 316 (and Ck.bar 506) is maintained at its original frequency, the bit error rate of a system that uses an ADC that incorporates latch circuit 600 can be improved.
FIG. 7 is a schematic diagram of a latch circuit 700, in which latch circuit 600 further comprises a second bistable pair 702. Preferably, second bistable pair 702 comprises a fifth PMOSFET “M10” 704 and a sixth PMOSFET “M11” 706. Ideally, M10 704 and M11 706 are matched transistors. Preferably, each of M10 704 and M11 706 has a gain greater than one. However, second bistable pair 702 can function if the product of the individual gains of M7 606 and M8 608 (i.e., the loop gain) is greater than one. The gate terminal of 706 is connected to the drain terminal of M10 704 and to the gate terminal of M8 608. The gate terminal of M10 704 is connected to the drain terminal of M11 706 and to the gate terminal of M5 408. The source terminals of M10 704 and M11 706 are together connected to second supply voltage VDD 404. The skilled artisan will appreciate that M10 704 and M11 706 can also be realized in other field effect, junction, or combination transistor technologies.
When the voltage of Ck.bar 506 is DOWN (i.e., the reset phase), M6 504 and M9 610 connect the gate and drain terminals of M10 704 and M11 706 together to second supply voltage VDD 404. With the gate and source terminals of M10 704 and M11 706 both connected to second supply voltage VDD 404, the gate-to-source voltages of M10 704 and M11 706 are made to equal zero, holding M10 704 and M11 706 OFF.
When the voltage of Ck.bar 506 is UP (i.e. the sampling phase), the states of M1 306, M2 308, M10 704, and M11 706 are controlled by first and second current signals i1 318 and i2 320. However, in the situation in which first and second current signals i1 318 and i2 320 both have values near to that of bias current ib (i.e., small signal current is), there can be a significant delay before first and second current signals i1 318 and i2 320 act to force one MOSFET (e.g., M1 306) OFF while the other MOSFET (e.g., M2 308) remains ON. Contemporaneously, with M4 406 and M7 606 both ON at the start of the sampling phase, the gate-to-source voltages of M5 408 and Mg 608 (i.e., PMOSFETs) can drift to values greater than their threshold voltages such that M5 408 and M8 608 turn ON. Having M1 308, M2 310, M4 406, M5 408, M7 606, and M8 608 all ON before the MOSFETs change states can cause latch circuit 700 to draw a large amount of current. Latch circuit 700 acts, in response to first and second current signals i1 318 and i2 320, to force one MOSFET of second bistable pair 702 (e.g., M10 704) ON while the other MOSFET of second bistable pair 702 (e.g., M11 706) remains OFF. The MOSFET of second bistable pair 702 (e.g., M10 704) that turns ON connects the gate terminal of its corresponding vertical latch MOSFET (e.g., M5 408) to second supply voltage VDD 404. With the gate and source terminals of the corresponding vertical latch MOSFET connected together, the gate-to-source voltage of the corresponding vertical latch MOSFET is made to equal zero, holding the corresponding vertical latch MOSFET OFF. In this manner, second bistable pair 702 acts to prevent latch circuit 700 from drawing unnecessary current before the MOSFETs change states during the sampling phase. Thus, for comparable realizations of latch circuits 600 and 700, latch circuit 700 consumes less power.
For example, when, at the start of the sampling phase, first current signal i1 318 is slightly larger than bias current ib (i.e., small positive signal current is), then first current signal ii 318 slowly continues to charge the parasitic capacitances at N4 310, which causes the voltage at N4 310 to rise slightly. This is indicated by a very small up-arrow “m” 708. Because the voltage at N4 310 is also the voltage at the gate terminal of M4 406, the voltage at the gate terminal of M4 406 rises by the same amount as the rise in the voltage at N4 310. This is indicated by a very small up-arrow “n” 710, where very small up-arrow n 710 has the same length (i.e., the same change in voltage) as very small up-arrow m 708.
Because the voltage at the source terminal of M4 406 is held at analog ground VAG 218, the gate-to-source voltage of M4 406 increases by the same amount as the rise in the voltage at the gate terminal of M4 406. The increase in the gate-to-source voltage of M4 406 causes its drain current to increase. In response to the increase in the gate-to-source voltage of M4 406 and the increase in its drain current, the drain-to-source voltage of M4 406 decreases by a greater magnitude than the increase in its gate-to-source voltage. This is indicated by a small down-arrow “p” 712, where small down-arrow p 712 has a longer length (i.e., a larger change in voltage) than very small up-arrow n 710. Because the voltage at the source terminal of M4 406 is held at analog ground VAG 218, the decrease in the drain-to-source voltage of M4 406 causes the voltage at the drain terminal of M4 406 to drop by the same amount.
Because the voltage at the drain terminal of M4 406 is also the voltage at the gate terminal of M10 704, the voltage at the gate terminal of M10 704 drops by the same amount as the drop in the voltage at the drain terminal of M4 406. This is indicated by a small down-arrow “q” 714, where small down-arrow q 714 has the same length (i.e., the same change in voltage) as small down-arrow p 712.
Because the voltage at the source terminal of M10 704 is held at second supply voltage VDD 404, the drop in the voltage at the gate terminal of M10 704 (i.e., a PMOSFET) causes its gate-to-source voltage to increase by the same amount. The increase in the gate-to-source voltage of M10 704 causes its drain current to increase. In response to the increase in the gate-to-source voltage of M10 704 and the increase in its drain current, the drain-to-source voltage of M10 704 decreases by a greater magnitude than the decrease in its gate-to-source voltage. This is indicated by a large up-arrow “r” 716, where large up-arrow r 716 has a greater length (i.e., a larger change in voltage) than small down arrow q 714. Because the voltage at the source terminal of M10 704 is held at second supply voltage VDD 404, the decrease in the drain-to-source voltage of M10 704 (i.e., a PMOSFET) causes the voltage at the drain terminal of M10 704 to rise by the same amount.
Because the voltage at the drain terminal of M10 704 is also the voltage at the gate terminal of M8 608, the voltage at the gate terminal of M8 608 rises by the same amount as the rise in the voltage at the drain terminal of M10 704. This is indicated by a large up-arrow “s” 718, where large up-arrow s 718 has the same length (i.e., the same change in voltage) as large up-arrow r 716. Because the voltage at the source terminal of M8 608 is held at second supply voltage VDD 404, the rise in the voltage at the gate terminal of M8 608 (i.e. a PMOSFET) causes its gate-to-source voltage to decrease by the same amount.
The decrease in the gate-to-source voltage of M8 608 ensures that it is less than its threshold voltage so that M8 608 is held OFF. Having M8 608 held OFF until first current signal i1 318 charges the parasitic capacitances at N4 310 to its new steady state voltage of HIGH prevents latch circuit 700 from drawing unnecessary current during the sampling phase.
Likewise, it can be demonstrated that when, at the start of the sampling phase, second current signal i2 320 is slightly larger than bias current ib (i.e., small negative signal current is), then second current signal i2 320 causes M11 706 to turn ON so that M5 408 is held OFF. Having M5 408 held OFF until second current signal i2 320 charges the parasitic capacitances at N5 312 to its new steady state voltage of HIGH prevents latch circuit 700 from drawing unnecessary current during the sampling phase.
Simulations of an implementation that uses an ADC that incorporates latch circuit 700, in which parameters that define latch circuit 700 (i.e., supply voltages, clock frequency, etc.) had specific values, showed latch circuit 700 to be capable of a five-fold increase in speed, or alternatively capable of reducing bit error rate from 10−10 to between 10−80 and 10−50.
FIG. 8 is a schematic diagram of a latch circuit 800, in which M3 314 of reset switch 304 of latch circuit 700 is replaced by a reset circuit 802. Preferably, reset circuit 802 comprises a sixth NMOSFET “M12” 804, a seventh NMOSFET “M13” 806, an eighth NMOSFET “M14” 808, and a ninth NMOSFET “M15” 810. The skilled artisan will appreciate that M12 804, M13 806, M14 808, and M15 810 can also be realized in other field effect, junction, or combination transistor technologies. In general, M14 808 and M15 810 can be realized in a variety of switch technologies, including microelectromechanical embodiments, while M12 804 and M13 806 can be realized using diodes.
In latch circuit 800, the source terminals of M1 306, M2 308, M4 406, and M7 606 are together connected to a third node “N6” 812. In reset circuit 802, the gate and drain terminals of M12 804 are together connected to N4 310. The gate and drain terminals of M13 806 are together connected to N5 312. The source terminals of M12 804 and M13 806 are together connected to the drain terminal of M14 808. The drain terminal of M15 810 is connected to N6 812. The source terminals of M14 808 and M15 810 are together connected to analog ground VAG 218. (Alternatively, analog ground VAG 218 can be replaced by first supply voltage “VSS”.) Clock waveform Ck 316 is applied to the gate terminal of M14 808. Inverse clock waveform Ck.bar 506 is applied to the gate terminal of M15 810.
With the gate and drain terminals of M12 804 connected together, M12 804 turns ON when its gate-to-source voltage is greater than its threshold voltage. Likewise, with the gate and drain terminals of M13 806 connected together, M13 806 turns ON when its gate-to-source voltage is greater than its threshold voltage. When the voltage of Ck 316 is UP (i.e., the reset phase), M14 808 reduces the voltages at the source terminals of M12 804 and M13 806 so that their gate-to-source voltages are greater than their threshold voltages and M12 804 and M13 806 turn ON. When M12 804 and M13 806 are ON, they connect N4 310 and N5 312 together. Contemporaneously, when the voltage of Ck.bar 506 is DOWN (i.e., the reset phase), M15 810 disconnects N6 812 from analog ground VAG 218. This insulates bistable pair 302, vertical latch 402, and second vertical latch 602 from the connection between N4 310 and N5 312 provided by M12 804 and M13 806. The connection between N4 310 and N5 312 provided by M12 804 and M13 806 causes the steady state voltages at both nodes to be equal. First and second current signals i1 318 and i2 320 flow through M12 804 and M13 806, which are sized so that, under these conditions, their drain-to-source voltages are MID. Thus, the voltages at N4 310 and N5 312 are equal to MID.
When the voltage of Ck.bar 506 is UP and the voltage of Ck 316 is DOWN (i.e., the sampling phase), M15 810 connects N6 812 to analog ground VAG 218, M14 808 raises the voltages at the source terminals of M12 804 and M13 806 so that their gate-to-source voltages are less than their threshold voltages and M12 804 and M13 806 turn OFF, and latch circuit 800 operates in the same manner as latch circuit 700 described above.
In latch circuits 300, 400, 500, 600, and 700, the source terminal of M3 314 is connected to the drain terminal of one of M1 306 or M2 308; the drain terminal of M3 314 is connected the drain terminal of the other of M1 306 or M2 308. Clock waveform Ck 316 is applied to the gate terminal of M3 314. When the voltage of Ck 316 is UP, M3 314 connects the gate and drain terminals of M1 306 and M2 308 together. This requires that the UP voltage of Ck 316 be greater than the sum of HIGH and the threshold voltage of M3 314. This can pose a problem when these latch circuits are realized in integrated circuits with low power supply voltages. Advantageously, in latch circuit 800, the source terminals of M14 808 and M15 810 are connected to analog ground VAG 218 so that the UP voltage of Ck 316 only needs to be greater than the threshold voltages of M14 808 and M15 810.
FIG. 9 shows a flow chart of a method 900 for decreasing the time in which a latch circuit port receiving a current signal greater than a bias current reaches a steady state voltage. The latch circuit comprises a bistable pair of cross connected transistors of a first polarity, and a third transistor of a second polarity. For example, the drain terminal of a first FET of a first polarity is connected to the gate terminal of a second FET of the first polarity, the drain terminal of the second FET is connected to the gate terminal of the first FET, the source terminals of the FETs are connected together, and a third FET of a second polarity is provided. In another example, the collector terminal of a first junction transistor of a first polarity is connected to the base terminal of a second junction transistor of the first polarity, the collector terminal of the second junction transistor is connected to the base terminal of the first junction transistor, the emitter terminals of the junction transistors are connected together, and a third junction transistor of a second polarity is provided.
At a step 902, the current signal greater than the bias current is amplified with the third transistor. At a step 904, the amplified current signal is applied to the latch circuit port receiving the current signal greater than the bias current. For example, a current signal greater than a bias current (e.g., i1 318) is received at a latch circuit port (e.g., N4 310) and continues to charge the parasitic capacitances at the latch circuit port, which causes the voltage at the latch circuit port to rise. The voltage at the latch circuit port is amplified by a third transistor (e.g., M5 408) so that it draws current from a power supply (e.g., VDD 404). The current drawn from the power supply is applied to the latch circuit port. Thus, the parasitic capacitances at the latch circuit port are charged under the combined effects of the current signal greater than the bias current and the current drawn from the power supply. This decreases the time in which the latch circuit port receiving the current signal greater than the bias current reaches the steady state voltage.
FIG. 10 is a schematic diagram of a latch circuit 1000, in which latch circuit 700 further comprises a reset circuit 1002. Preferably, reset circuit 1002 comprises a tenth NMOSFET “M16” 1004 and an eleventh NMOSFET “M17” 1006. The skilled artisan will appreciate that M16 1004 and M17 1006 can also be realized in other field effect, junction, or combination transistor technologies. In general, M16 1004 and M17 1006 can be realized in a variety of switch technologies, including microelectromechanical embodiments.
In latch circuit 1000, the source terminals of M1 308 and M2 310 are together connected to third node N6 812, and the source terminals of M4 406 and M7 606 are together connected to a fourth node “N7” 1008. In reset circuit 1002, the drain terminal of M161004 is connected to N6 812, and the drain terminal of M17 1006 is connected to N7 1008. The source terminals of M16 1004 and M17 1006 are together connected to analog ground VAG 218. (Alternatively, analog ground VAG 218 can be replaced by first supply voltage “VSS”.) Clock waveform Ck 316 is applied to the gate terminal of M16 1004. Inverse clock waveform Ck.bar 506 is applied to the gate terminal of M17 1006.
When the voltage of Ck 316 is UP and the voltage of Ck.bar 506 is DOWN (i.e., the reset phase), M16 1004 connects N6 812 to analog ground VAG 218, and M17 1006 disconnects N7 1008 from analog ground VAG 218. With the voltages at the source terminals of M4 406 and M7 606 not held equal to analog ground VAG 218, the gate-to-source voltages of M4 406 and M7 606 are less than their threshold voltages so that M4 406 and M7 606 turn OFF. Having M4 406 and M7 606 turned OFF prevents them from drawing current during the reset phase. Thus, for comparable realizations of latch circuits 700 and 1000, latch circuit 1000 consumes less power.
When the voltage of Ck.bar 506 is UP and the voltage of Ck 316 is DOWN (i.e., the sampling phase), M16 1004 disconnects N6 812 from analog ground VAG 218, and M17 1006 connects N7 1008 to analog ground VAG 218, and latch circuit 1000 operates in the same manner as latch circuit 700 described above.
FIG. 11 is a schematic diagram of a latch circuit 1100, in which latch circuit 400 further comprises reset circuit 1002. In latch circuit 1100, the source terminals of M1 308 and M2 310 are together connected to third node N6 812, and the source terminal of M4 406 is connected to fourth node N7 1008. When the voltage of Ck 316 is UP and the voltage of Ck.bar 506 is DOWN (i.e., the reset phase), M16 1004 connects N6 812 to analog ground VAG 218, M17 1006 disconnects N7 1008 from analog ground VAG 218, and reset circuit 1102 operates in the same manner as described above. When the voltage of Ck.bar 506 is UP and the voltage of Ck 316 is DOWN (i.e., the sampling phase), M16 1004 disconnects N6 812 from analog ground VAG 218, and M17 1006 connects N7 1008 to analog ground VAG 218, and latch circuit 1100 operates in the same manner as latch circuit 400 described above.
FIG. 12 is a schematic diagram of a latch circuit 1200, in which latch circuit 500 further comprises a voltage source “ΔV1” 1202. Voltage source ΔV1 1202 is connected between the source terminal of M6 504 and second supply voltage VDD 404.
When the voltage of Ck.bar 506 is DOWN (i.e., the reset phase), M6 504 connects the gate terminal of M5 408 to voltage source ΔV1 1202. Voltage source ΔV1 1202 is set to a voltage level that holds M5 408 OFF when the gate terminal of M5 408 is connected to voltage source ΔV1 1202. This disrupts the latching action of vertical latch 402 so that bistable pair 302 can assume a state independent of the state of vertical latch 402 as described above.
However, unlike latch circuit 500, which holds the voltage at the gate terminal of M5 408 equal to the voltage of second supply voltage VDD 404, latch circuit 1200 holds the voltage at the gate terminal of M5 408 equal to the difference between the voltages of supply voltage VDD 404 and voltage source ΔV1 1202. This reduces the time in which M6 504 changes state from ON to OFF when the voltage of Ck.bar 506 changes from DOWN to UP (i.e., the sampling phase). In turn, this reduces the time in which vertical latch 402 can resume its latching action to decrease the time necessary for the port (i.e., N4 310) receiving the current signal (i.e., i1 318) that is greater than bias current ib to reach its new steady state voltage. The skilled artisan will appreciate that the function of voltage source ΔV1 1202 could be realized using a variety of devices including, but not limited to, a resistor, a diode-connected MOSFET, or a bias current source.
FIG. 13 is a schematic diagram of a latch circuit 1300, in which latch circuit 700 further comprises voltage source ΔV1 1202 and a second voltage source “ΔV2” 1302. Voltage source ΔV1 1202 is connected between the source terminal of M6 504 and second supply voltage VDD 404. Second voltage source ΔV2 1302 is connected between the source terminal of M9 610 and second supply voltage VDD 404.
Second voltage source ΔV2 1302 operates in the same manner as voltage source ΔV1 1202. Whereas, when the voltage of Ck.bar 506 changes from DOWN to UP (i.e., the sampling phase), voltage source ΔV1 1202 reduces the time in which M6 504 changes state from ON to OFF, second voltage source ΔV2 1302 reduces the time in which M9 610 changes state from ON to OFF. Because the time in which either port (i.e., N4 310 or N5 312) receiving current signal (i.e., i1 318 or i2 320) that is greater than bias current ib reaches its new steady state voltage is decreased, the frequency of Ck 316 (and Ck.bar 506) can be increased. This can increase the processing speed of an ADC that incorporates latch circuit 1300. Alternatively, the bit error rate of a system that uses an ADC that incorporates latch circuit 1300 can be improved. As is the case with voltage source ΔV1 1202, the function of second voltage source ΔV2 1302 could be realized using a variety of devices.
FIG. 14 is a schematic diagram of a latch circuit 1400, in which vertical latch reset switch 502 and second vertical latch reset switch 604 of latch circuit 700 are replaced by a third vertical latch reset switch 1402. Preferably, third vertical latch reset switch 1402 comprises a seventh PMOSFET “M18” 1404. The skilled artisan will appreciate that M18 1404 can also be realized in other field effect, junction, or combination transistor technologies. In general, M18 1404 can be realized in a variety of switch technologies, including microelectromechanical embodiments.
In latch circuit 1400, the source terminals of M5 408, M8 608, M10 704, and M11 706 are together connected to a fifth node “N8” 1406. In third vertical latch reset switch 1402, the drain terminal of M18 1404 is connected to N8 1406. The source terminal of M18 1404 is connected to second supply voltage VDD 404. Clock waveform Ck 316 is applied to the gate terminal of M18 1404.
Latch circuit 1400 is configured so that, when the voltage of Ck 316 is DOWN (i.e., the sampling phase), the gate-to-source voltage of M181404 (i.e., a PMOSFET) is less than its threshold voltage and M18 1404 turns ON. When M18 1404 is ON, it connects N8 1406 to second supply voltage VDD 404, and latch circuit 1400 operates in a similar manner as latch circuit 700 described above.
However, when the voltage of Ck 316 is UP (i.e. the reset phase), the gate-to-source voltage of M18 1404 (i.e., a PMOSFET) is greater than its threshold voltage so that M18 1404 turns OFF. This reduces the voltage at the source terminals of M5 408, M8 608, M10 704, and M11 706 (i.e., PMOSFETs) so that their gate-to-source voltages are less than their threshold voltages and M5 408, M8 608, M10 704, and M11 706 also turn OFF. In turn, this reduces the voltages at the drain terminals of M4 406 and M7 606 (i.e., NMOSFETs) so that their drain-to-source voltages are less than the differences between their gate-to-source voltages and their threshold voltages, and M4 406 and M7 606 also turn OFF. Having M4 406 and M7 606 turned OFF prevents them from drawing current during the reset phase. Thus, for comparable realizations of latch circuits 700 and 1400, latch circuit 1400 consumes less power.
FIG. 15 is a graph 1500 of a current 1502 drawn by latch circuit 700 as a function of time “t” 1504 after the voltage of Ck 316 changes from UP to DOWN. Graph 1500 comprises a series of curves including, but not limited to, a first curve 1506, a second curve 1508, a third curve 1510, and a fourth curve 1512.
At steady state, latch circuit 700 draws current IvDD 1502 equal to four times bias current ib. Recalling Eqs. (1) and (2), the sum of the currents drawn by M1 306 and M2 308 is equal to twice bias current ib. Additionally, M4 406 mirrors the current drawn by M2 308, while M7 606 mirrors the current drawn by M1 306. (During the sampling phase, the currents drawn by M4 406 and M7 606 are from the parasitic capacitances associated with, respectively, M5 408 and M8 608.)
In latch circuit 700, during the reset phase, M1 306, M2 308, M4 406, and M7 606 are turned ON, while M5 408, M8 608, M10 704, and M11 706 are turned OFF. During the sampling phase, the states of M1 306 and M2 308 are controlled by first and second current signals i1 318 and i2 320. For example, if first current signal i1 318 is greater than bias current ib, M1 306 turns OFF while M2 308 remains ON. Under these conditions, at steady state, M5 408 and M10 704 also turn ON, while M7 606 turns OFF. M4 406 remains ON, and M8 608 and M11 706 remain OFF. However, although M5 408 and M10 704 are ON, because M1 306 and M7 606 are OFF, latch circuit 700 does not draw any additional current IvDD 1502. In graph 1500, this situation is indicated by first curve 1506, which equals a steady state current of four times bias current ib.
As mentioned above, the time in which M1 306 and M2 308 change states is a function of the sizes of first and second current signals i1 318 and i2 320. If the current signal (e.g., i1 318 or i2 320) received at the port (e.g., N4 310 or N5 312) of the MOSFET changing state from ON to OFF is sufficiently large, the port reaches its new steady state voltage relatively quickly, and the appropriate MOSFETs of latch circuit 700 also change states relatively quickly. Thus, during a relatively quick transient, again no additional current IvDD 1502 is drawn by latch circuit 700. This situation is indicated by second curve 1508, which also equals a steady state current of four times bias current ib.
However, if the time of the transient becomes longer, latch circuit 700 acts to decrease the time needed for the port (e.g., N4 310 or N5 312) receiving the current signal (e.g., i1 318 or i2 320) greater than bias current ib to reach its new steady state voltage. In this situation, latch circuit 700 draws current IvDD 1502 as charted by, for example, third curve 1510. As indicated by the shape of third curve 1510, current IvDD 1502 drawn by latch circuit 700 increases at a relatively slow rate, reaches a peak value 1514, then decreases relatively quickly. Peak value 1514 is reached when the latching action of latch circuit 700 occurs.
If the time of the transient becomes increasingly longer, latch circuit 700 draws current IvDD 1502 for a longer period of time, as charted by, for example, fourth curve 1512. As indicated by the shape of fourth curve 1512, current IvDD 1502 drawn by latch circuit 700 increases at the same rate as indicated by third curve 1510. However, current IvDD 1502 is drawn for a longer period of time resulting in fourth curve 1512 having a higher peak value 1516.
Graph 1500 shows how latch circuit 700 decreases the time needed for the port (e.g., N4 310 or N5 312) receiving the current signal (e.g., i1 318 or i2 320) greater than bias current ib to reach its new steady state voltage, while limiting the power consumed to realize this decrease in time. Latch circuit 700 only draws current IvDD 1502 in those situations in which first and second current signals i1 318 and i2 320 both have values near to that of bias current ib (i.e., small signal current is).
Of note, each curve of graph 1500 is separated from its next curve by an equal separation in time “Δt” 1518. However, each curve represents a ten-fold magnitude increase in current signal (e.g., i1 318 or i2 320) over the next curve to the right. For example, current signal (e.g., i1 318 or i2 320) for curve 1510 is one-thousand times greater than current signal (e.g., i1 318 or i2 320) for curve 1512.
FIG. 16 is a schematic diagram of a comparator 1600 implemented using latch circuit 700. Comparator 1600 comprises an input stage 1602, a latch circuit 1604, and an output stage 1606. Preferably, input stage 1602 comprises a differential amplifier 1608, a first current mirror 1610, and a second current mirror 1612.
Preferably, differential amplifier 1608 comprises a differential pair 1614 and a current source 1616. Preferably, differential pair 1614 comprises amplifying MOSFETs “M19” 1618 and “M20” 1620. Preferably, current source 1616 comprises biasing MOSFETs “M21” 1622 and “M22” 1624. A biasing MOSFET is connected to the source terminal of each amplifying MOSFET. The drain terminal of “M21” 1622 is connected to the source terminal of M19 1618; the drain terminal of “M22” 1624 is connected to the source terminal of M20 1620. The drain terminals of M21 1622 and M22 1624 are also connected together. The source terminals of M21 1622 and M22 1624 are together connected to analog ground VAG 218. (Alternatively, analog ground VAG 218 can be replaced by first supply voltage “VSS”.) A load MOSFET is connected to the drain terminal of each amplifying MOSFET. The drain terminal of “M23” 1626 is connected to the drain terminal of M19 1618; the drain terminal of “M24” 1628 is connected to the drain terminal of M20 1620. The source terminals of M23 1626 and M24 1628 are together connected to second supply voltage VDD 404.
Preferably, first current mirror 1610 comprises a MOSFET “M25” 1630, and second current mirror 1612 comprises a MOSFET “M26” 1632. The source terminals of M25 1630 and M26 1632 are together connected to second supply voltage VDD 404. The gate terminal of M25 1630 is connected to the gate and drain terminals of M23 1626; the gate terminal of M26 1632 is connected to the gate and drain terminals of M24 1628. In input stage 1602, M19 1618, M20 1620, M21 1622, and M22 1624 are NMOSFETs, while M23 1626, M24 1628, M25 1630, and M26 1632 are PMOSFETs. However, this configuration can be reversed depending upon the overall configuration of comparator 1600.
The voltage of analog signal x 228 is received by input stage 1602 at a first input port “P1” 1634, which is the noninverting terminal of comparator 1600 (e.g., A 202, B 204, or C 206). This allows the voltage of analog signal x 228 to be compared with the reference voltage received at second input port “P2” 1636, which is the inverting terminal of comparator 1600. For example, the voltage of analog signal x 228 is compared with V/4, for comparator A 202; V/2, for comparator B 204; and 3V/4, for comparator C 206. First input port Pi 1634 is connected to the gate terminals of M191618 and M21 1622. Second input port P2 1636 is connected to the gate terminals of M20 1620 and M22 1624.
Differential pair 1614 (i.e., M19 1618 and M20 1620) acts to control the distribution of current flowing through current source 1616 (e.g., M21 1622 and M22 1624). The sum of the current flowing through both M19 1618 and M20 1620 equals the current provided by current source 1616. For example, as the voltage received at first input port Pi 1634 rises with respect to the voltage received at second input port P2 1636, the portion of the total current that flows through M19 1618 and M23 1626 increases, while the portion of the total current that flows through M20 1620 and M24 1628 decreases. M25 1630 mirrors the increase in current flowing through M23 1626 to produce first current signal i1 318 at the drain terminal of M25 1630. M26 1632 mirrors the decrease in current flowing through M24 1628 to produce second current signal i2 320 at the drain terminal of M261632.
In the above explanation, differential amplifier 1608 is configured so that the voltage of analog signal x 228 provides bias for M21 1622 of current source 1616, while the reference voltage provides bias for M22 1624 of current source 1616. The skilled artisan will appreciate that differential amplifier 1608 can also be configured with a traditional current source that is independently biased (i.e., the bias is not provided by the voltage of analog signal x 228 or the reference voltage).
Latch circuit 1604 comprises latch circuit 700, a third current mirror 1638, and a fourth current mirror 1640. Preferably, third current mirror 1638 comprises a MOSFET “M27” 1642, and fourth current mirror 1640 comprises a MOSFET “M28” 1644. The source terminals of M27 1642 and M28 1644 are together connected to second supply voltage VDD 404. The gate terminal of M27 1642 is connected to the gate terminal of M5 408; the gate terminal of M28 1644 is connected to the gate terminal of M8 608. In comparator 1600, latch circuit 700 can be replaced by any of latch circuits 800, 1000, 1300, or 1400. In latch circuit 1604, M27 1642 and M28 1644 are PMOSFETs. However, depending upon the overall configuration of comparator 1600, they can be NMOSFETs.
First current signal i1 318 and second current signal i2 320 are received at, respectively, N4 310 and N5 312 and are processed by latch circuit 700 as described above. M27 1642 mirrors the current flowing through M5 408 to produce a third current signal “i3” 1646 at the drain terminal of M27 1642. M28 1644 mirrors the current flowing through M8 608 to produce a fourth current signal “i4” 1648 at the drain terminal of M28 1644. Third current signal i3 1646 and fourth current signal i4 1648 are proportional to, respectively, the voltages at N4 310 and N5 312 (i.e., either HIGHER or LOWER).
Preferably, output stage 1606 comprises a hold latch 1650. Preferably, hold latch 1650 comprises a third bistable pair 1652 and a fourth bistable pair 1654. Preferably, third bistable pair 1652 comprises a first MOSFET “M29” 1656 and a second MOSFET “M30” 1658. The gate terminal of M30 1658 is connected to the drain terminal of M29 1656 at a first port “N9” 1660. The gate terminal of M29 1656 is connected to the drain terminal of M30 1658 at a second port “N10” 1662. The source terminals of M29 1656 and M30 1658 are together connected to second supply voltage VDD 404. Preferably, fourth bistable pair 1654 comprises a third MOSFET “M31” 1664 and a fourth MOSFET “M32” 1666. The gate terminal of M32 1666 is connected to the drain terminal of M31 1664 at first port N9 1660. The gate terminal of M31 1664 is connected to the drain terminal of M32 1666 at second port N10 1662. The source terminals of M31 1664 and M32 1666 are together connected to analog ground VAG 218. (Alternatively, analog ground VAG 218 can be replaced by first supply voltage “VSS”.) In output stage 1606, M29 1656 and M30 1658 are PMOSFETs, while M31 1664 and M32 1666 are NMOSFETs. However, this configuration can be reversed depending upon the overall configuration of comparator 1600.
Third current signal i3 1646 is received at N9 1660, while fourth current signal i4 1648 is received at N10 1662. Typically, output stage 1606 is followed by digital logic circuits. Sometimes when latch circuit 1604 is resetting, the voltages at N4 310 and N5 312 are neither HIGHER nor LOWER, but some value in between. This can cause problems in the digital logic circuits. Hold latch 1650 retains the output of latch circuit 1604 prior to reset, which is either HIGHER or LOWER. Thus, output stage 1606 serves as a buffer between latch circuit 1604 and the digital logic circuits.
Another advantage of vertical latch circuit 402 can be observed by comparing it with hold latch 1650. In hold latch 1650, the sum of the gate-to-source voltage of M29 1656 and the gate-to-source voltage of M31 1664 is equal to the difference between VDD 404 and VAG 218. Implementations of hold latch 1650 must take this relationship into consideration. In contrast, for the same values of VDD 404 and VAG 218, each of M4 406 and M5 408 can realize a gate-to-source voltage that is larger than that of M29 1656 or M31 1664. Alternatively, by using vertical latch 402, the difference between VDD 404 and VAG 218 can be reduced, which can reduce the power consumed.
FIG. 17 is a schematic diagram of an alternative embodiment 1700 of vertical latch 402. Preferably, vertical latch 1700 comprises a first current minor pair 1702 and a second current mirror pair 1704. Preferably, first current minor pair 1702 comprises NMOSFET M4 406 and a twelfth NMOSFET “M33” 1706. Preferably second current mirror pair 1704 comprises PMOSFET M5 408 and an eighth PMOSFET “M34” 1708. The skilled artisan will appreciate that M33 1706 and M34 1708 can also be realized in other field effect, junction, or combination transistor technologies. M33 1706 and M34 1708 can also be realized using diodes.
In first current minor pair 1702, the drain and gate terminals of M33 1706 are connected together, the gate terminal of M33 1706 is connected to the gate terminal of M4 406, the drain terminal of M33 1706 is connected to the drain terminal of M5 408, and the source terminals of M33 1706 and M4 406 are together connected to analog ground VAG 218. In second current pair 1704, the drain and gate terminals of M34 1708 are connected together, the gate terminal of M34 1708 is connected to the gate terminal of M5 408, the drain terminal of M34 1708 is connected to the drain terminal of M4 406, and the source terminals of M34 1708 and M5 408 are together connected to second supply voltage VDD 404.
M4 406, M5 408, M33 1706, and M34 1708 are sized so that the product of the current gain of first current mirror pair 1702 and the current gain of second current minor pair 1704 (i.e., the loop gain of vertical latch 1700) is greater than one. This ensures that vertical latch 1700 will have a latching action. Vertical latch 1700 can also be an alternative embodiment for second vertical latch 602, with M4 406 replaced by M7 606, and M5 408 replaced by M8 608.
Diode-connected M33 1706 and M34 1708 provide vertical latch 1700 with several advantages. They provide bias voltages for, respectively, M4 406 and M5 408. This is particularly important for M5 408, which, absent vertical latch reset switch 502, lacks a bias voltage necessary to operate during the sampling phase.
Furthermore, the skilled artisan will appreciate that including vertical latch 402 in latch circuit 400 complicates problems with controlling the dynamic offset voltages in the latch circuit. These problems are particularly troublesome in vertical latch 402 because it comprises both NMOSFET M4 406 and PMOSFET M5 408, and therefore entails the difficulties associated with matching different MOSFET types. (In latch circuit 600, the difficulties arise in matching corresponding MOSFETs (i.e., M4 406 with M7 606, and M5 408 with M3 608).) However, diode-connected M33 1706 and M34 1708 can be sized to bias, respectively, M4 406 and M5 408 in a manner that corrects the detrimental effects of their offset voltages.
Additionally, while each of the various latch circuit configurations presented above is designed to decrease the time necessary for the port (i.e., N4 310 or N5 312) receiving the current signal (i.e., i1 318 or i2 320) that is greater than bias current ib to reach its new steady state voltage, several of the configurations are also designed to limit the power consumed while realizing this decrease in time. Diode-connected M33 1706 and M34 1708 enable a designer to bias, respectively, M4 406 and M5 408 in a manner that controls when they will change states during a transient. This allows the designer to balance the competing needs for decreasing the time necessary for the latch circuit to reach steady state and limiting the power consumed by the latch circuit.
FIG. 18 shows a flow chart of a method 1800 for reducing the power consumed by a latch circuit. The latch circuit comprises a bistable pair and a vertical latch, wherein the bistable pair has a first MOSFET (e.g., M1 306) and a second MOSFET (e.g., M2 308) configured so that the drain terminal of the first MOSFET is connected to the gate terminal of the second MOSFET at a first port (e.g., N4 310), the drain terminal of the second MOSFET is connected to the gate terminal of the first MOSFET at a second port (e.g., N5 312), and the source terminals of the first and second MOSFETs are connected together, and wherein the vertical latch has a third MOSFET (e.g., M4 406) and a fourth MOSFET (e.g., M5 408) configured so that the gate terminal of the third MOSFET is connected to the gate terminal of the second MOSFET and the drain terminal of the fourth MOSFET, and the gate terminal of the fourth MOSFET is connected to the drain terminal of the third MOSFET.
At a step 1802, the bistable pair and vertical latch are reset. In one alternative, at a step 1804, the fourth MOSFET is held OFF during the resetting, thereby reducing the power consumed by the latch circuit during the resetting. For example, the voltage at the source terminal of the fourth MOSFET (e.g., M5 408) is held constant, while the voltage at the gate terminal of the fourth MOSFET is changed, so that the fourth MOSFET is held OFF during the resetting. In another example, the voltage at the drain terminal of the fourth MOSFET (e.g., M5 408) is held constant, while the voltage at the source terminal of the fourth MOSFET is changed, so that the fourth MOSFET is held OFF during the resetting.
In another alternative, at a step 1806, the third MOSFET is held OFF during the resetting, thereby reducing the power consumed by the latch circuit during the resetting. For example, the voltage at the gate terminal of the third MOSFET (e.g., M4 406) is held constant, while the voltage at the source terminal of the third MOSFET is changed, so that the third MOSFET is held OFF during the resetting. In another example, the voltage at the source terminal of the third MOSFET (e.g., M4 406) is held constant, while the voltage at the drain terminal of the third MOSFET is changed, so that the third MOSFET is held OFF during the resetting.
In yet another alternative, at a step 1808, after the resetting, the fourth MOSFET is held OFF when the second MOSFET changes state from ON to OFF, thereby reducing the power consumed by the latch circuit after the resetting. For example, after the resetting, the voltage at the source terminal of the fourth MOSFET (e.g., M5 408) is held constant, while the voltage at the gate terminal of the fourth MOSFET is changed, so that the fourth MOSFET is held OFF when the second MOSFET changes state from ON to OFF.
Conclusion
While various embodiments of the present invention have been described above, it should be understood that they have been presented by way of example, and not limitation. It will be apparent to persons skilled in the relevant art that various changes in form and detail can be made therein without departing from the spirit and scope of the invention. Thus the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents.

## Claims

1. A latch circuit, comprising:
a bistable latch that includes:
a first bistable latch transistor and a second bistable latch transistor;
a first port to receive a first input current signal and produce a first output voltage signal; and
a second port to receive a second input current signal and produce a second output voltage signal; and

a second latch separate from the bistable latch and connected to the first port of the bistable latch to provide a feedback signal to the first port based on the first output voltage signal, the second latch including:
a first latch transistor having a source terminal configured to directly connect to a power supply voltage; and
a second latch transistor having a source terminal directly connected to a ground potential,

wherein the first port reaches a steady state value based on the first and second input current signals and the feedback signal,
wherein a source of the first bistable latch transistor is directly connected to the ground potential, and
wherein a source of the second bistable latch transistor is directly connected to the ground potential.

2. The latch circuit of claim 1, wherein the feedback signal is a feedback voltage signal.

3. The latch circuit of claim 1, wherein the feedback signal is a feedback current signal.

4. The latch circuit of claim 3, wherein the feedback current signal grows exponentially.

5. The latch circuit of claim 1, further comprising a latch reset switch connected to the second latch to disrupt the latching action of the second latch and reduce power consumption of the second latch during a reset phase.

6. The latch circuit of claim 5, further comprising a reset voltage supply associated with the latch reset switch to reduce the time in which the second latch resumes its latching action.

7. The latch circuit of claim 1, further comprising a third latch connected to the second port of the bistable latch to receive the second output voltage signal and provide a second feedback signal to the second port based on the second output voltage signal.

8. The latch circuit of claim 1, wherein the first and second output voltages reach steady state values based on parasitic capacitances.

9. The latch circuit of claim 1, wherein the second latch enables at least one of: an increased clock frequency and an improved bit error rate (BER) associated with the latch circuit.

10. An analog-to-digital converter, comprising:
a comparator having a first input to receive an analog signal and a second input to receive a reference signal, the comparator producing a digital signal and including a latch circuit, the latch circuit including:
a bistable latch that includes:
a first bistable latch transistor and a second bistable latch transistor;
a first port to receive a first input current signal and produce a first output voltage signal; and
a second port to receive a second input current signal and produce a second output voltage signal; and

a second latch separate from the bistable latch and connected to the first port of the bistable latch to provide a feedback signal to the first port based On the first output voltage signal, the second latch including:
a first latch transistor having a source terminal configured to directly connect to a power supply voltage; and
a second latch transistor having a source terminal directly connected to a ground potential,

wherein the first port reaches a steady state value based on tie first and second input current signals and the feedback signal,
wherein a source of the first bistable latch transistor is directly connected to the ground potential, and
wherein a source of the second bistable latch transistor is directly connected to the ground potential.

11. The analog-to-digital converter of claim 10, wherein the feedback signal is a feedback voltage signal.

12. The analog-to-digital converter of claim 10, wherein the feedback signal is a feedback current signal.

13. The analog-to-digital converter of claim 12, wherein the feedback current signal grows exponentially.

14. The analog-to-digital converter of claim 10, further comprising a latch reset switch connected to the second latch to disrupt the latching action of the second latch and reduce power consumption of the second latch during a reset phase.

15. The analog-to-digital converter of claim 14, further comprising a reset voltage supply associated with the latch reset switch to reduce the time in which the second latch resumes its latching action.

16. The analog-to-digital converter of claim 10, further comprising a third latch connected to the second port to receive the second output voltage signal and provide a second feedback signal to the second port based on the second output voltage signal.

17. The analog-to-digital converter of claim 10, wherein the first and second output voltages reach steady state values based on parasitic capacitances.

18. The analog-to-digital converter of claim 10, wherein the second latch enables at least one of: an increased clock frequency and an improved bit error rate (BER) associated with the latch circuit.

19. A flash analog-to-digital converter, comprising:
a voltage ladder to provide a plurality of reference voltages;
an input stage to produce a differential current signal associated with each of the plurality of reference voltages and an input analog signal, each differential current signal including first and second input current signals; and
a plurality of latch comparators each receiving the first and second input current signals and producing a digital output signal, each latch comparator including:
a bistable latch that includes:
a first bistable latch transistor and a second bistable latch transistor;
a first port to receive the first input current signal and produce a first output voltage signal; and
a second port to receive a second input current signal and produce a second output voltage signal; and

a second latch separate from the bistable latch and connected to the first port of the bistable latch to provide a feedback signal to the first port based on the first output voltage signal, the second latch including:
a first latch transistor having a source terminal configured to directly connect to a power supply voltage; and
a second latch transistor having a source terminal directly connected to a ground potential;

wherein the first port reaches a steady state value based on the first and second input current signals and the feedback signal,
wherein a source of the first bistable latch transistor is directly connected to the ground potential, and
wherein a source of the second bistable latch transistor is directly connected to the ground potential.

20. The flash analog-to-digital converter of claim 19, wherein the second latch enables at least one of: an increased clock frequency and an improved bit error rate (BER) associated with the corresponding latch comparator.

21. A latch circuit, comprising:
a bistable latch that includes:
a first port to receive a first input current signal and produce a first output voltage signal; and
a second port to receive a second input current signal and produce a second output voltage signal; and
a first bistable latch transistor having a gate terminal and a source terminal; and
a second bistable latch transistor having a gate terminal and a source terminal,
the source terminal of the first bistable latch transistor being directly connected to the source terminal of the second bistable latch transistor, each of the source terminal of the first bistable latch transistor and the source terminal of the second bistable latch transistor being directly connected to the ground potential,
the gate terminal of the first bistable latch transistor being directly connected to the second port, and
the gate terminal of the second bistable latch transistor being directly connected to the first port; and

a second latch separate from the bistable latch and connected to the first port of the bistable latch to provide a feedback signal to the first port based on the first output voltage signal, the second latch including:
a first latch transistor having a source terminal configured to directly connect to a power supply voltage; and
a second latch transistor having a source terminal directly connected to a ground potential,

wherein the first port reaches a steady state value based on the first and second input current signals and the feedback signal.

22. The latch circuit of claim 6, the reset voltage supply being connected between the power supply voltage and the latch reset switch.