# Patent

## Title
Low-voltage comparator-based switched-capacitor networks

## Abstract

Described is a switched-capacitor network and method for performing an analog circuit function. The circuit includes a switched-capacitor network, a comparator, and a voltage-offset network. The switched-capacitor network includes multiple switches, each having a respective threshold voltage and connected to one of a high-limit voltage, a low-limit voltage, and electrical ground. A first comparator input terminal in communication with the switched-capacitor network is configured to receive a node voltage therefrom during a first phase. The second input terminal is configured to receive one of the high-limit voltage and the low-limit voltage. The voltage-offset network provides a voltage shift at the first input terminal setting an input reference level at a mid-level voltage with respect to the high-limit voltage and the low-limit voltage. The voltage shift enables the first terminal to receive full-swing voltages when the high-limit voltage is less than twice the threshold voltage, with power supply voltages below twice the threshold voltage.

## Background
FIELD OF THE INVENTION
The present invention relates generally to switched-capacitor circuits and, more particularly, to low-voltage switched-capacitor circuits for comparator-based integrated circuits.
BACKGROUND OF THE INVENTION
Modern scaled complementary metal-oxide semiconductor (CMOS) processes are typically optimized for digital circuits. Process advancements such as lower voltage power supplies and shorter gate lengths result in low power, high-speed digital circuits, but can also result in higher power, low performance analog circuits. Lower output resistance, reduced power supply voltage, increased threshold variation and gate leakage present design challenges for analog and mixed signal systems.
The design of high-gain operational amplifiers (hereinafter op-amps) is one example of a design challenge resulting from the continued scaling of CMOS processes. High gain op-amps are critical components of many analog and mixed-signal circuits, and are especially important in switched-capacitor implementations of analog circuits including integrators, filters and other applications including analog-to-digital converters. As gate length decreases, the intrinsic gain per unit current of a device also decreases. Although a smaller gate length increases the transconductance gm, the reduction in the output resistance r0 dominates. Moreover, it is not practical to maintain an acceptable intrinsic gain per unit current by using longer devices in a scaled implementation, especially when increased frequency capability is required. In addition, the output resistance of modern scaled devices is not linearly proportional to gate length; increasing the gate length does not significantly increase the output resistance of the device.
Scaled processes generally utilize lower voltages to prevent gate oxide damage or device breakdown during operation. To achieve satisfactory gain in an amplifier designed in a scaled process, it is often necessary to utilize a cascode topology; however, a cascode topology using a reduced supply voltage generally results in a substantially reduced voltage swing. Modern low-voltage scaled processes result in inherently less gain and voltage swing than older processes, consequently widely used analog design styles such as switched-capacitor networks need to be modified to compensate for these effects. Switched-capacitor circuits demand high performance from op-amps included in the circuits. In a highly scaled CMOS process it is generally difficult to achieve the required op-amp performance.
SUMMARY OF THE INVENTION
The present invention addresses the design challenges associated with the use of high-gain op-amps within switched-capacitor circuits, by incorporating a comparator-based architecture in place of the high-gain op-amps. The comparator-based switched-capacitor circuits can be combined with low-voltage techniques to enable operation at supply levels approaching a single transistor gate threshold voltage.
In one aspect, the invention features a switched-capacitor network for performing an analog circuit function. The circuit includes a switched-capacitor network having an input terminal to receive an input voltage and multiple switches, each switch has a respective threshold voltage. Each switch is in communication with one of a high-limit voltage, a low-limit voltage, and electrical ground. The circuit also includes a comparator having an output terminal, a first input terminal, and a second input terminal. The first input terminal is in communication with the switched-capacitor network and is configured to receive a node voltage from the switched-capacitor network during a first phase for sampling the input voltage. The second input terminal is configured to receive one of a high-limit voltage and a low-limit voltage. The circuit also includes a voltage-offset network in communication with the first input terminal. The voltage-offset network provides a voltage shift at the first input terminal, setting an input reference level at a mid-level voltage with respect to the high-limit and low-limit voltages. Also included in the circuit is a controllable current source coupled between the output terminal and one of the high-limit and low-limit voltages. The controllable current source has a control input coupled to the output terminal of the comparator. A current is supplied by the controllable current source during a second phase, sweeping the output voltage toward the other one of the high-limit and low-limit voltages.
In another aspect, the invention features a method for performing an analog circuit function in a circuit that includes a comparator in communication with a switched-capacitance network. An input voltage is sampled by the switched-capacitance network during a first phase. The switched-capacitance network includes multiple switches each having a respective threshold voltage and in communication with one of a high-limit voltage, a low-limit voltage, and electrical ground. A voltage present at a node within the switched-capacitance network is applied to a first comparator input terminal. One of the high-limit voltage and the low-limit voltage is applied to a second comparator input terminal, and a voltage shift is applied to the first comparator input terminal. The applied voltage shift sets an input reference level at a mid-level voltage with respect to the high-limit voltage and the low-limit voltage.



BRIEF DESCRIPTION OF THE DRAWINGS
The above and further advantages of this invention may be better understood by referring to the following description in conjunction with the accompanying drawings, in which like numerals indicate like structural elements and features in the various figures. The drawings are not necessarily to scale, emphasis instead being placed upon illustrating the principles of the invention.
FIG. 1 is a circuit diagram of a conventional switched-capacitor integrator.
FIG. 2A is a circuit diagram of an embodiment of a comparator-based switched-capacitor network in accordance with the present invention.
FIG. 2B is a circuit diagram of an alternative embodiment of the comparator-based switched-capacitor network shown in FIG. 2A.
FIG. 2C is a circuit diagram of another alternative embodiment of the comparator-based switched-capacitor network shown in FIG. 2A.
FIG. 3 is an exemplary timing diagram showing non-overlapping clock signals used to control the switches in the circuit of FIG. 2.
FIG. 4A is a circuit diagram depicting the effective circuit of FIG. 2 during a sampling phase.
FIG. 4B is a circuit diagram depicting the effective circuit of FIG. 2 during a reset phase.
FIG. 4C is a circuit diagram depicting the effective circuit of FIG. 2 during an evaluation phase.
FIG. 5 is a graphical representation of the output voltage as a function of time for the circuit of FIG. 2.
FIG. 6 is a more detailed circuit diagram of an embodiment of a multi-stage comparator-based switched-capacitor network in accordance with the present invention.
FIG. 7 is a circuit diagram of an embodiment of a differential comparator-based switched-capacitor network in accordance with the present invention.
FIG. 8 is a circuit diagram of an embodiment of an optional common-mode feedback network that can be applied to the differential circuit of FIG. 7 in accordance with the present invention.



DETAILED DESCRIPTION
In brief overview, the present invention relates to a comparator-based switched-capacitor network for performing an analog circuit function under low-voltage conditions. The circuit includes a switched-capacitor network, a comparator, and a voltage-offset network. Other techniques are used to enable operation at low voltages. Fore example, in a CMOS implementation, individual switches of the switched-capacitor network are implemented as PMOS or NMOS-only transistors. Each switch is coupled to one of three reference values: a high-limit voltage, a low-limit voltage, or an electrical ground potential. The use of any series-connected switches that are not coupled to one of these reference values is avoided. The high-limit and low-limit voltages are close to power supply rail voltages. This leaves some headroom to avoid inadvertently turning on any of the switches or forward biasing any junction due to an overshoot condition.
An input voltage is sampled by the switched-capacitor network during a first phase. A node voltage of the switched-capacitor network reflective of the sampled input voltage is applied to a first input terminal of the comparator. A second comparator input terminal providing a reference level is coupled to the low-limit voltage. The voltage-offset network provides a voltage shift at the first input terminal setting an input reference level at a mid-level voltage with respect to the high-limit and low-limit voltages. The voltage shift enables the first terminal to receive full-swing voltages when the high-limit voltage is less than twice the threshold voltage.
A second switched-capacitor network is connected between the first comparator input terminal and circuit output terminal, providing a circuit output voltage thereon. A reset circuit temporarily pulls the output voltage to one of the high-limit and low-limit voltages during a reset phase. A controllable current source is coupled to the circuit output terminal with its control input coupled to the comparator output terminal. The current source transfers charge from the second switched-capacitor network, driving the output voltage towards an opposite one of the limiting voltages during a second phase. The resulting charge transfer also alters the potential at the first comparator input terminal through the second switched capacitor network. At a “correct” value, the comparator output changes state causing the current source to turn off. After the current source is turned off, the output voltage remains substantially constant until at least the conclusion of the second phase. The resulting output voltage level generally depends upon charge previously stored within the second switched capacitor network, the sampled input voltage, and component values.
The resulting circuit can be implemented as an integrator and used as a basic building block for a variety of analog and mixed-signal circuit applications, including filters and analog-digital converters.
A conventional switched-capacitor integrator 10 is shown in FIG. 1. The integrator 10 includes an op-amp 12 having an output terminal 18, a non-inverting input terminal 14, and an inverting input terminal 16. The op-amp's output terminal 18 is coupled to a circuit output terminal 20. A feedback capacitor Cint1 is coupled between the output terminal 18 and the inverting input terminal 14. The non-inverting input terminal 16 is coupled to a ground reference potential. A switched-capacitor network 22 is coupled between the inverting terminal 14 and a circuit input terminal 24.
The switched-capacitor network 22 includes an input capacitor CIN1 coupled at a first terminal to the input terminal 24 through a first series-connected switch S11 and at a second terminal to the inverting terminal 14 through a second series-connected switch S14. A first shunt-connected switch S12 is connected between the first terminal of the input capacitor CIN1 and electrical ground. A second shunt-connected switch S13 is similarly connected between the second terminal of the input capacitor CIN1 and electrical ground. A reference node VX1 is defined at the interconnection of the second terminal of the input capacitor CIN1, the second shunt-connected switch S13 and the second series-connected switch S14.
Circuit operation is controlled by two non-overlapping clock phases: an input, or a sampling phase φ1 and an evaluation phase φ2. The clock phases are used to control the switches S11, S12, S13, S14 between on and off (i.e., short-circuited and open-circuited) states. Different sets of clock phases applied to the switches result in different integrator transfer functions. As one example, a clock phasing that results in stray-insensitive inverting integration is now described. Each of the switches S11, S12, S13, S14 is marked with a respective one of the two phases φ1, φ2, indicating which of the phases is used to control the switch. In particular, during the sampling phase φ1, the first series connected switch S11 and the second shunt-connected switch S13 are closed; whereas, the second series-connected switch S14 and the first shunt-connected switch S12 are opened. In this configuration, the input voltage VIN1 is applied to the first terminal of the input capacitor CIN1, depositing a charge onto the input capacitor CIN1 resulting in a sampling of input signal VIN onto the input capacitor CIN1.
During the evaluation phase φ2, the first series-connected switch S11 and the second shunt-connected switch S13 are opened; whereas, the second series-connected switch S14 and the first shunt-connected switch S12 are closed. In this configuration, at least a portion of the charge stored on the input capacitor CIN1 results in a node voltage VX, that disturbs the virtual ground condition when coupled to the inverting terminal 14. The inverting terminal 14 can also be referred to as a virtual ground terminal 14 in that it resides at the same potential as the non-inverting terminal 16 (i.e., ground) under stead-state conditions. In this configuration, the op-amp 12 is also connected in a negative feedback topology, with its output terminal 18 connected to the virtual ground terminal 14 through the feedback capacitor Cint1. Through such a connection the output of the op-amp 12 drives the voltage at its inverting input terminal 14 until it equals the voltage at its non-inverting input terminal 16 in steady-state. In this manner, the op-amp forces the virtual ground node VX1 to a virtual ground potential during the second phase φ2, thereby redistributing charge between the input capacitor CIN1 and the feedback capacitor Cint1.
Through the principle of charge conservation, charge originally stored on the input capacitor CIN1 has effectively been moved to the feedback, or integrating capacitor Cint1, resulting in a change in output voltage ΔVOUT1 at the circuit output terminal 20 as described in Equation 1.

ΔVOUT1=VIN1*[CIN1/Cint1]  (1)

Because the change in output voltage ΔVOUT1 is proportional to the input voltage VIN1, this circuit 10 functions as an integrator. Note that the op-amp 12 in this switched-capacitor system 10 consumes static power to maintain the virtual ground node voltage at all times.
All switches S11, S12, S13, and S14 are connected either to electrical ground or virtual ground with the exception of the input series switch S11. For low-voltage conditions in which the supply voltage (VDD) approaches a transistor threshold voltage, a CMOS transmission gate configuration of the series switch S11 would be unable pass mid-level voltages. Such a series switch S11 coupled to the input terminal 24 therefore limits the useful input voltage range under low-voltage conditions. Unable to pass mid-level voltages, the circuit 10 is unable to accommodate full-swing input voltage VIN1 without distortion.
An approach referred to as a “switched-op-amp technique” uses circuit topologies that avoid problems associated with passing full-swing voltage signals through series connected switches. In switched-op-amp topologies, each of the switches S11, S12, S13, and S14 operates under a respective constant voltage level near a power supply rail voltage (e.g., VDD, VSS, or ground). Consequently, each of the switches S11, S12, S13, and S14 can be implemented in either PMOS-only or NMOS-only, without the need for having a transmission gate configuration. Whether a switch is PMOS or NMOS generally depends whether the switch is connected to VDD or ground. For example, NMOS switches can be used in connecting to VSS or ground; whereas, PMOS switches can be used to connect to VDD. Without CMOS transmission gate switches, signal distortion is avoided under low-voltage operation. Additionally, a DC operating point of the virtual ground node is also chosen to be close to one of the power supply rails (VDD, VSS, or ground), rather than the middle voltage, commonly referred to as the common-mode voltage.
To accommodate full-scale input voltages VIN1, the input series-connected switch S11 is eliminated altogether. In the case of cascaded integrators (multiple stages connected in an output terminal 20 to input terminal 24 fashion), the function of the removed series-connected switch S11 is performed by enabling/disabling the output of the preceding integrator. Disabling the output of the preceding integrator can be accomplished by switching off the amplifier that drives that output. Because no full-swing voltage signals are passed through any of the remaining switches S12, S13, S14, switched-op-amp topologies can work at power supply voltages as low as VDD,min≈Vt where Vt represents the larger of Vtn (the NMOS threshold voltage) or |Vtp| (the absolute value of the PMOS threshold voltage).
One difficulty associated with the switched-op-amp topology occurs in the input of the first integrator stage, where there is no preceding op-amp to turn off. More generally, there is no guarantee about what kind of circuitry provides the input voltage VIN1. Thus, circuitry is designed to allow for a voltage difference between the output terminal of the previous stage and the input terminal 10 of the first integrator stage.
In some embodiments, the integrator input 24 is protected by a series-connected input resistor RIN. Such a resistor RIN limits the maximum current that can be drawn by the input terminal 24. Unfortunately, such an input resistor RIN low-pass filters the input signal by the combination of the input resistor RIN and the input capacitor CIN1. To minimize any undesirable effects, these component values should be chosen such that the resulting bandwidth of the R-C filter is substantially greater than the anticipated input signal bandwidth.
In some embodiments, the value of the series-connected input resistor RIN is chosen as a compromise between the effective off and on output resistance values ROFF, RON of the replaced series-connected switch S11. The series-connected input resistor RIN is chosen to be large enough to avoid shorting the output of the previous stage during the evaluation phase φ2. The series-connected input resistor RIN is also chosen to be small enough to prevent the resulting R-C low-pass filter from excessively filtering the input signal. It should be noted that similar to their switched-capacitor counterparts, switched-op-amp topologies also require high-gain op-amps, with the added requirement that these op-amps include an enable/disable feature.
In accordance with the principles of the present invention, the switched-op-amp technique is combined with comparator-based architectures resulting in relaxed requirements for comparator-based switched-capacitor topologies. One possible application of the combined technique has been demonstrated in the form of a low-voltage comparator-based switched-capacitor integrator.
FIG. 2A is a circuit diagram of a first embodiment of a comparator-based switched-capacitor integrator 30 in accordance with the present invention. The integrator 30 includes a comparator 32 having an output terminal 33, an inverting input terminal 34, and a non-inverting input terminal 36. A first switched-capacitor network 42 is coupled between an integrator input terminal 38 and the comparator's inverting terminal 34. A second switched-capacitor network 46 is coupled between an integrator output terminal 40 and inverting comparator input terminal 34. The non-inverting input terminal 36 is further coupled to a low-limit source Vref,LO, close to one of the power supply rails (e.g., VSS) rather than the ground reference potential of the op-amp circuit 10 (FIG. 1).
The first switched-capacitor network 42 includes a series-connected input capacitor CIN2 coupled at one end to the input terminal 38 through a series-connected input resistor RIN2 and at another end to the comparator's inverting terminal 34. The series-connected input resistor RIN2 is provided in a first input stage circuit in place of a series-connected input switch S11 (FIG. 1). A first shunt-connected switch S21 is connected between the input terminal of the input capacitor CIN2 and a high-limit source Vref,HI. A second shunt-connected switch S22 is connected between the output terminal of the input capacitor CIN2 and the low-limit source Vref,LO. A reference node VX2 is defined between the first switched capacitor network 42, the comparator's inverting input terminal 34, the second shunt-connected switch S22 and a series-connected feedback switch S25.
The values of the high and low-limit sources Vref,HI, Vref,LO are close to, but not equal to the power supply rail voltages (VDD, VSS), allowing for some overshoot of the output signal. Without some allowance for overshoot, the voltage at node VX2 might exceed the power supply rails during output voltage sweeping because of comparator delay. This overshoot would forward bias the normally reverse biased PN junctions of the transistor switches, leading to inaccuracy through loss of charge.
The second switched-capacitor network 46 includes a feedback capacitor Cint2 coupled between the circuit output terminal 40 and one end of the series-connected feedback switch S25. One end of the feedback capacitor Cint2 is also coupled to the high-limit source Vref,HI through a series-connected reset switch S26. The other end of the feedback switch S25 is coupled to the comparator's inverting input terminal 34. In the exemplary embodiment, the circuit output terminal 40 is also connected to the low-limit source Vref,LO through a current source 48. In other embodiments, the current source could source current from a different voltage than Vref,LO. The current source 48 providing a substantially constant current ID2 is controllable by an output signal VCOMP2 provided at the comparator's output terminal 33.
An offset voltage network 44 is coupled between each of the high-limit and low-sources Vref,HI, Vref,LO and the reference node VX2. The offset voltage network 44 includes an offset capacitor CIN2/2 coupled at one end to the reference node VX2. The other end of the offset capacitor CIN2/2 is coupled to the low-limit source Vref,LO through a first series-connected switch S23 and to the high-limit source Vref,HI through a second series-connected switch S24. As indicated, in some embodiments the capacitance of the offset capacitor is approximately half the value of the input capacitor CIN2 to produce a desired effect of setting an input reference voltage at mid-level with respect to the high-limit and low-limit sources Vref,HI, Vref,LO.
Circuit operation is controlled by three timing phases: an input, or sampling phase φ1; a non-overlapping output, or evaluation phase φ2; and a brief reset phase φR. Thus, the phases of the integration cycle are: InputResetOutput.
FIG. 3 is an exemplary timing diagram showing clock signals used to control the switches of the comparator-based switched-capacitor integrator 30. The input phase φ1 is ON for a sample period and OFF elsewhere. The output phase φ2 is ON during an evaluation period and OFF elsewhere. The input and output phases φ1, φ2 do not overlap. In some embodiments, a brief reset phase φR overlaps an initial portion of the output phase φ2.
FIG. 4A is a circuit diagram depicting the effective circuit of FIG. 2A during the input phase φ1. With the input phase φ1 ON, the second shunt-connected switch S22 of the first switched-capacitor network 42 and the second series-connected switch S24 of the offset voltage network 44 are closed and have been replaced by short circuits. With the output phase φ2 OFF the first shunt-connected switch S21, the second series connected switch S24 and the series-connected feedback switch S25 are open and have been replaced by open circuits. For the exemplary embodiment in which the reset phase φR overlaps a portion of the output phase φ2, the reset switch S26 is also OFF and has been replaced by an open circuit.
During the input phase φ1, the input capacitor CIN2 is coupled between the input voltage VIN2 and the low-limit source Vref,LO through the series input resistor RIN2. Consequently, an input-dependent charge, ΔQX, is deposited onto the summing node VX2. Likewise, the offset capacitor CIN2/2 is coupled between Vref,HI and Vref,LO and charged to a preset value.
During the output phase φ2, the charge ΔQX deposited on the summing node VX2 is transferred to the feedback capacitor Cint2 by manipulating the output voltage, VOUT2 until the summing node voltage VX2 is restored back to virtual ground.
FIG. 4B is a circuit diagram depicting the effective circuit of FIG. 2A during a reset phase φR. With the reset signal φR ON, the reset switch S26 of the second switched-capacitor network 46 is closed and has been replaced by a short circuit. Because the reset phase φR overlaps an initial portion of the output phase φ2, the first shunt-connected switch S21 of the first switched-capacitor network 42, the first series-connected switch S23 of the offset voltage network 44, and the series-connected feedback switch of the second switched-capacitor network 46 are also closed and have been replaced by short circuits. With the first timing signal φ1 OFF the second shunt-connected switch S22 and the second series connected switch S24 are open and have been replaced by open circuits.
In this configuration, the input side of the previously charged input capacitor CIN2 is pulled up to the high-limit source Vref,HI; the previously charged voltage offset capacitor CIN2/2 is pulled down to the low-limit source Vref,LO; and the output 40 is coupled to the comparator input terminal 34 through the feedback capacitor Cint2.
During the reset phase φR, the circuit output terminal 40 is shorted (i.e., reset) to one of the high-limit and low-limit sources Vref,HI, Vref,LO. In the exemplary embodiment, the output terminal 40 is shorted to the high-limit source, Vref,HI. In other embodiments, the output terminal 46 can be shorted to the low-limit source Vref,LO. For applications, such as the exemplary integrator 30, it is also important to preserve any charge previously stored on the integrating capacitor Cint2. With the switch configuration described above, any stored charge will be distributed among all of the coupled capacitors Cint2, CIN2, CIN2/2. The redistribution of charge results in a change of the potential of the summing node VX2, altering the voltage relationship between the two input terminals 34, 36 and causing the output state VCOMP of the comparator 32 to “flip.” For example, the output of the comparator changes from a HIGH state to a LOW state. The output state VCOMP controls operation of the current source 48. For example, a LOW state turns the current source 48 on, such that a substantially constant current value ID2 flows in the direction indicated from the output terminal 40 (from the integration capacitor Cint2) to the low limit source Vref,LO, effectively discharging the output 40.
At the end of the reset phase φR, the reset switch S26 opens and is replaced by an open circuit. All other switches remain unchanged. FIG. 4C is a circuit diagram depicting the effective circuit of FIG. 2A during an evaluation phase. During the output, or evaluation phase, the current source 48 sweeps the output voltage VOUT2 linearly towards the opposite limit source thereby discharging the output voltage VOUT2. When the virtual ground node VX2 reaches a threshold of the comparator, nominally equal to the voltage Vref, LO applied to the non-inverting input 36, the comparator output VCOMP2 changes state again. As the comparator output state VCOMP2 had been LOW state, it transitions to a HIGH state. This transition controls the current source 48, effectively turning off the current source 48. The output voltage VOUT2 is then held constant for at least the remainder of the evaluation phase φ2. A slower (second) controllable current source 51 can be used to reverse the overshoot that is caused by comparator delay as shown in FIG. 2C.
The central idea behind comparator-based switched-capacitor designed is that the output of the sampled system need only be correct at the instant the sample of the output voltage is taken at the end of the evaluation phase. In a switched-capacitor integrator 30, this means that the voltage of the virtual ground node VX2 need only be correct at the sampling instant defined by the instant the comparator output VCOMP2 flips. As long as this condition is met, there is no need to guarantee the potential of the virtual ground node VX2 at other times. This results in a power savings improvement over traditional op-amp circuits that consume power to maintain the virtual ground node at a virtual ground potential.
FIG. 5 is a graphical representation of the output voltage VOUT2(t) as a function of time for the circuit of FIG. 2A. The output voltage VOUT2(t) is shown for at least three samples: n, n+1, and n+2. For the n+1st sample, the output voltage VOUT2(t) is initially held at the previous sampled value VOUT(n) during a first phase φ1. During a reset phase φR, the output voltage is pulled up to the high-limit source Vref,HI. At the conclusion of the reset phase φR, the current source linearly discharges the voltage of the output node VOUT2, until the comparator output changes state, which turns off the current source ID2.
When the output voltage reaches the correct value, the virtual ground node will simultaneously be at the desired value, analogous to the traditional switched-capacitor approach using op-amps. At this point, the comparator output flips polarities, shutting off the current source, preserving the correct output voltage. The output voltage VOUT(n+1) remains constant at least until the conclusion of the output phase φ2. The particular output voltage VOUT(n+1) depends upon the previously stored value and the current input value. In general the comparator-based switched-capacitor implementation is applicable to all switched-capacitor networks such as A/D converters, delta-sigma modulators, amplifiers, and filters. Also note that the comparator-based switched-capacitor approach is inherently superior to the switched-op-amp technique, because all current sources connected to the output are switched off after the output phase eliminating the need to switch the op-amp on or off. There is no need to disable the comparator 32 as there had been for the op-amp 12 (FIG. 1).
The capacitor of size CIN/2 is used in the voltage-offset network to create a DC voltage shift at the input terminal 38. This voltage shift maximizes input signal swing by setting the input referenced level to the midlevel voltage, (Vref,HI+Vref,LO)/2. After one complete integration cycle, the change in charge ΔQX2 is provided in Equation 2.

ΔQX2=ΔVX2CX,tot=(Vref,HI−Vref,LO)CIN2+(Vref,LO−Vref,HI)CIN2/2+ΔVOUT2Cint2  (2)

The change in voltage at the summing node is provided in Equation 3.

ΔVX2=ΔQX2/CX,tot  (3)

Where CX,tot is the sum of all capacitance is at node VX2. The total capacitance CX,tot is given in Equation 4.

CX,tot=CIN2+CIN2/2+Cint2  (4)

Because the comparator stops discharging the output node when ΔVX2=0, therefore ΔQX=0 and Equation 2 can be solved for the change in voltage at the output, ΔVOUT2.

ΔVOUT2=CIN2/Cint2(VIN2−(Vref,LO+Vref,HI)/2)  (5)

The output changes by the input value measured with respect to the mid-level voltage, with a gain determined by the capacitance ratio. This is the desired function for a discreet time integrator.
One detail of the proposed circuit is that the reference voltages, Vref,LO and Vref,HI, are used instead of using a VDD and ground directly from the power supply. In some embodiments, the reference voltages are derived from a band-gap voltage source. It should be noted that at least one band-gap derived reference voltage is normally required in any accurate analog-to-digital converter (ADC) because using both power supply rails and signal references would allow power supply noise to leak into the system.
It is expected that the use of a second voltage reference will not affect the minimum allowed power supply voltage, VDD,min. In order to turn on switches connected to reference voltages, VDD,min≈Vt−|Vrail−Vref|, which means this design does not reach the ideal switched op-amp VDD,min of Vt. However, even in the op-amp-based design, VDD cannot practically be so low because of a sub-threshold leakage current.
In another embodiment, the output voltage VOUT2 is coupled to a fourth switched capacitor network 49 as shown in FIG. 2B. The switched capacitor network 49 comprises at least one input capacitor CIN3 and a sampling switch S27. The operating principle of the illustrated embodiment is identical to that of the embodiment of FIG. 2A except the comparator output VCOMP2 controls the state of the switch S27 instead of the current source ID2. The sampling switch S27 turns ON when VCOMP2 is LOW and turns off when VCOMP2 is HIGH. As in FIG. 2A, when the virtual ground node VX2 reaches a threshold of the comparator 32 during the evaluation period, nominally equal to the voltage Vref,LO applied to the non-inverting input 36, the comparator output VCOMP2 changes state again. As the comparator output state VCOMP2 had been in a LOW state, it transitions to a HIGH state. This transition controls the switch S27, effectively sampling the voltage VOUT2 across the input capacitor CIN3.
FIG. 6 is a more detailed circuit diagram of another embodiment of the invention illustrating a multi-stage comparator-based switched-capacitor network 50. The multi-stage circuit 50 includes at least two stages 62, 64, substantially similar to the comparator-based switched-capacitor integrator 30 (FIG. 2). Additionally, each of the switches has been implemented in one of an NMOS or a PMOS transistor. Generally, a PMOS transistor switch (e.g., switches M1a, M4a, M6a, M1b, M4b, M6b) is used when connecting to the high-limit voltage Vref, HI and an NMOS transistor switch (e.g., switches M2a, M3a, M5a, M2b, M3b, M5b) is used when connecting to the low-limit source Vref,LO. In this exemplary embodiment, the input to the first stage 62 is preceded by a series resistor RIN3, as used in the switched op-amp technique. Each of the switches of the first stage 62 is similarly controlled by a respective one of the phases φ1, φ2, φR2, with φR2 being the reset phase that partially overlaps φ2. In the second stage 62, the first and second phases φ1, φ2 have been reversed. Also, the reset phase φR1 is used, which overlaps phase φ1 instead of φ2. Thus, in the first stage 62, switches M2a, M4a are controlled by the first phase φ1; whereas, in the second stage 64, corresponding switches M2b, M4b are controlled by the second phase φ2.
Likewise, in the first stage 62, switches M1a, M3a, and M5a are controlled by the second phase φ2; whereas, in the second stage 64, corresponding switches M1b, M3b, and M5b are controlled by the first phase φ2. The output voltage VOUT3a of the first stage 62 is equivalent to the input voltage VIN3b of the second stage 64. No series input resistor is necessary for the second stage 64, since the output current sources of the first stage 62 are turned off after the completion of the evaluation phase φ2. Additional stages can be cascaded in a similar manner, with the phases similarly reversed in an alternating manner between adjacent stages.
Alternatively, the comparator output VCOMP3a controls the state of the switch M2b instead of controlling the current source 68, in a manner analogous to the embodiment illustrated in FIG. 2B.
The exemplary embodiments described thus far have been configured for single-ended signaling applications. In other embodiments, the comparator-based switched-capacitor networks are adapted for fully differential signaling applications. FIG. 7 is a circuit diagram of an embodiment of a differential comparator-based switched-capacitor integrator circuit 80. The differential circuit 80 includes a positive path 81 and a negative path 83. Each of the paths 81, 83 includes a respective input terminal 96, 98, with a differential input signal being applied between the two input terminals 96, 98. Each of the paths 81, 83 can optionally be preceded by a series input resistor RIN4p, RIN4n as described above for single-ended applications.
The positive path 81 is coupled to an inverting input terminal 84 of a differential comparator 82. Similarly, the negative path 83 is coupled to a non-inverting input terminal 86 of the same differential comparator 82. The positive path 81 includes a first switched-capacitor network including a first shunt-connected switch S41p connected to a high-limit source Vref,HI, second shunt-connected switch S42p connected to a low-limit source Vref,LO, with a series capacitor CIN4p connected therebetween. The negative path 83 includes a similar network. The positive path 81 also includes a voltage-offset network including a first series-connected switch S43p connected to the high-limit source Vref,HI and a second series-connected switch S44p connected to the low-limit source Vref,LO, with one end of each switch S43p, S43n further connected to one end of a series connected capacitor CIN4p/2. Once again, the negative path 83 includes a similar voltage-offset network.
Each path 81, 83 also includes a respective second switched-capacitor network. Each of the second switched-capacitor networks includes a respective feedback capacitor Cint4p, Cint4n and series-connected feedback switch S45p, S45n coupled between the respective output terminal 100, 102 and a respective virtual ground node VX4p, VX4n. The second switched-capacitor network of the positive path 81 includes a reset switch S46p connecting the positive output terminal 100 to the high voltage reference Vref,HI during the brief reset phase, φR. Similarly, the second switched-capacitor network of the negative path 83 includes a reset switch S46n connecting the negative output terminal 102 to the low-limit source Vref,LO during the same brief reset phase, φR. In general, during the reset phase, the outputs are reset to opposite references. With the reset phase overlapping an initial portion of the evaluation phase φ2, the feedback capacitors Cint4p, Cint4n are connected by the closed switches S45p, S45n to their respective differential virtual ground nodes VX3p, VX4n.
During the evaluation phase φ2, the output voltages VOUT4p, VOUT4n are then swept in opposite directions by the respective current source ID4p, ID4n, which allows the output common-mode voltage to remain at mid-level at all times during the integration cycle.
The fully differential comparator will include an offset voltage source VOS 104 at one of the input terminals 84 with a voltage value equal to Vref,HI−Vref,LO. The offset voltage source 104 is required because the positive virtual ground node VX4p has a nominal voltage of Vref,LO, while the negative virtual ground node VX4n has a nominal voltage of Vref,HI.
When the output has the correct value, VXn−VXp=Vref,HI−Vref,LO, the differential comparator output state changes and the current sources ID4p, ID4n are turned off. The output voltages VOUT4p, VOUT4n remain substantially constant for at least the remainder of the integration phase. Ignoring common-mode adjustments, VXp≈Vref,LO and VXn≈Vref,HI.
Preferably, applications using the low-voltage comparator-based switched-capacitor network operate as fast as possible. The comparator only needs to be very accurate for input voltages close to Vref,HI (plus some margin for common-mode adjustments). It will most likely consist of a number of cascaded gains stages.
In some embodiments, one or more similar differential stages can be cascaded with the input of one stage being taken from the output of a preceding stage. As described above in relation to FIG. 6, the first and second clock phases φ1, φ2 would be inverted in an alternating manner between adjacent stages. In alternative embodiments, the comparator outputs VCOMP4sp and VCOMP4an control sampling switches of additional switched-capacitor networks in a manner similar to the embodiment shown in FIG. 2B.
FIG. 8 illustrates one embodiment of a common-mode feedback circuit 110 that can be incorporated into the differential integrator 80 (FIG. 7) to control the output common-mode voltage. The common-mode feedback circuit 110 includes a first input 112 coupled to a common-mode reference node VCM through a first common-mode feedback capacitor CCMFBp. The input end of the capacitor CCMFBp is coupled to the high-limit voltage Vref,HI through a series connected switch S81 controlled by the reset clock phase φR. The common-mode feedback circuit 110 also includes a second input 114 coupled to the common-mode reference node VCM through a second common-mode feedback capacitor CCMFBn. The input end of the capacitor CCMFBn is coupled to the low-limit source Vref,LO through a series connected switch S82 also controlled by the reset clock phase φR.
The common-mode reference node VCM is connected to a bias potential Vbiasp through a series connected switch S83, also controlled by the reset clock phase φR and to a control terminal 115 (i.e., gate) of a transistor M81. Vbiasp controls the nominal value of offset current Icharge. Another terminal of the transistor 116 (i.e., the source) is coupled to the positive power supply rail, VDD. The series-connected switch S84 is controlled by a signal determined as the product (i.e., AND) of the second input clock phase φ2 and a control signal Vctrl. Vctrl is derived from the output of the comparator 82 (FIG. 7) and is used to turn on and off the current source transistor M81. The control terminal 115 is further coupled to the positive power supply rail voltage VDD through a second series-connected switch S85. The second series-connected switch S85 is controlled by a signal determined as the OR combination of the first clock phase φ1 and the complement of voltage control signal Vctrl. A third terminal of the transistor M81 is coupled to the respective charging discharging current source ID4p, ID4n, supplying an offset current Icharge to counter a drift in the common mode voltage level.
Two common-mode feedback capacitors CCMFBp, CCMFBn create a voltage divider between the positive and negative integrator outputs 102, 104 such that node VCM tracks the integrator's output common-mode voltage. During the reset phase, the voltage across these capacitors CCMFBp, CCMFBn is set such that the operating point of the node VCM is equal to the desired gate voltage of the current source transistor M81 to balance ID4p, ID4n. As the potential of the node VCM drifts away from its operating point voltage, the gate voltage of current source transistor M81 is adjusted to adjust the current Icharge that counters this common-mode drift. Although only one is necessary, two of these common-mode feedback circuits 110 can be used: one for the positive path 81 and one for the negative path 83 of the differential integrator 80.
Low-voltage comparator-based switched-capacitor networks can be used as building blocks in a variety of analog and mixed-signal applications, such as filters and analog-digital converters. For example, the low-voltage comparator-based switched-capacitor networks can be used to implement a sigma-delta analog to digital converter (ADC), such as the sigma-delta ADC described in U.S. patent application Ser. No. 11/343,064, filed Jan. 30, 2006, entitled “Comparator-Based Switched Capacitor Circuit for Scaled Semiconductor Fabrication Processes,” the entirety of which is incorporated herein by reference. In some embodiments, such an ADC can provide 12 to 14 bits of resolution with an input bandwidth of 200 kHz to 3 MHz.
While the invention has been shown and described with reference to specific embodiments, it should be understood by those skilled in the art that various changes in form and detail may be made therein without departing from the spirit and scope of the invention. For example, the descriptions above are directed to an integrator; however, the switched-capacitor network of the present invention can be utilized in other circuit embodiments such as sample-and-hold circuits, analog-to-digital converters and filters.

## Claims

1. A network for performing an analog circuit function comprising:
a first switched-capacitor network having an input terminal to receive an input voltage and a plurality of switches, each switch having a respective threshold voltage and in communication with one of a high-limit voltage, a low-limit voltage, and electrical ground;
a comparator having a comparator output terminal, a first comparator input terminal, and a second comparator input terminal, the first comparator input terminal in communication with the first switched-capacitor network and configured to receive a node voltage therefrom during a first phase for sampling of the input voltage, the second comparator input terminal configured to receive one of the high-limit voltage and the low-limit voltage, the comparator providing a first voltage or a second voltage at the comparator output terminal according to whether a voltage at the first comparator input terminal exceeds a voltage at the second comparator input terminal;
a voltage-offset network in communication with the first comparator input terminal, the voltage-offset network providing a voltage shift at the first comparator input terminal setting an input reference level at a mid-level voltage with respect to the high-limit and low-limit voltages;
a first controllable current source coupled to a network output terminal and having a control terminal coupled to the comparator output terminal, the first controllable current source supplying a current during a second phase to sweep a network output voltage toward one of the high-limit and low-limit voltages;
a second switched-capacitor network coupled at one end to the first comparator input terminal and at another end to the network output terminal;
a reset circuit coupled to the network output terminal and charging the second switched-capacitor network to the other one of the high-limit voltage and the low-limit voltage between the first phase and the second phase; and
a second controllable current source having a control terminal in communication with the network output terminal and supplying a current to compensate for a voltage error generated by a finite delay in a response of the comparator.

2. The network of claim 1, wherein each switch of the plurality of switches is selected from the group consisting of NMOS transistor and PMOS transistors.

3. The network of claim 1, wherein the switched-capacitor network is implemented in CMOS.

4. The network of claim 1, wherein the current source comprises a transistor.

5. The network of claim 1, wherein the second switched-capacitor network includes a feedback capacitor, the analog circuit function comprising integration of the input voltage.

6. The network of claim 1, further comprising a series-connected resistor receiving the input voltage at the input terminal of the first switched-capacitor network.

7. The network of claim 1, further comprising a substantially similar second stage having an input terminal in communication with the network output terminal.

8. The network of claim 7, further comprising a common-mode correction network coupled to the network output terminal and injecting a current thereto derived from a common-mode signal.

9. The network of claim 1, wherein the input voltage is obtained from a differential input signal and wherein the first and second switched-capacitor networks, the comparator, and the voltage-offset network are adapted for differential signal operation.

10. The network of claim 1, wherein the comparator is a non-clocked comparator.

11. A network for performing an analog circuit function comprising:
a first switched-capacitor network having an input terminal to receive an input voltage and a plurality of switches, each switch having a respective threshold voltage and in communication with one of a high-limit voltage, a low-limit voltage, and electrical ground;
a comparator having a comparator output terminal, a first comparator input terminal, and a second comparator input terminal, the first comparator input terminal in communication with the first switched-capacitor network and configured to receive a node voltage therefrom during a first phase for sampling of the input voltage, the second comparator input terminal configured to receive one of the high-limit voltage and the low-limit voltage, the comparator providing a first voltage or a second voltage at the comparator output terminal according to whether a voltage at the first comparator input terminal exceeds a voltage at the second comparator input terminal;
a voltage-offset network in communication with the first comparator input terminal, the voltage-offset network providing a voltage shift at the first comparator input terminal setting an input reference level at a mid-level voltage with respect to the high-limit and low-limit voltages;
a sampling capacitor coupled to a network output terminal;
a sampling switch coupled between the sampling capacitor and the one of the high-limit and low-limit voltages, the sampling switch having a control terminal coupled to the comparator output terminal;
a controllable current source coupled to the network output terminal and having a control terminal coupled to the comparator output terminal, the controllable current source supplying a current during a second phase to sweep a network output voltage toward one of the high-limit and low-limit voltages;
a second switched-capacitor network coupled at one end to the first comparator input terminal and at another end to the network output terminal; and
a reset circuit coupled to the network output terminal and charging the second switched-capacitor network to the other one of the high-limit and low-limit voltages between the first phase and the second phase.

12. The network of claim 11, further comprising a second controllable current source having a control terminal in communication with the network output terminal and supplying a current to compensate for a voltage error generated by a finite delay in a response of the comparator.

13. The network of claim 11, wherein each switch of the plurality of switches is selected from the group consisting of NMOS transistor and PMOS transistors.

14. The network of claim 11, wherein the switched-capacitor network is implemented in CMOS.

15. The network of claim 11, wherein the current source comprises a transistor.

16. The network of claim 11, wherein the second switched-capacitor network includes a feedback capacitor, the analog circuit function comprising integration of the input voltage.

17. The network of claim 11, further comprising a series-connected resistor receiving the input voltage at the input terminal of the first switched-capacitor network.

18. The network of claim 11, further comprising a substantially similar second stage having an input terminal in communication with the network output terminal.

19. The network of claim 18, further comprising a common-mode correction network coupled to the network output terminal and injecting a current thereto derived from a common-mode signal.

20. The network of claim 11, wherein the input voltage is obtained from a differential input signal and wherein the first and second switched-capacitor networks, the comparator, and the voltage-offset network are adapted for differential signal operation.

21. The network of claim 11, wherein the comparator is a non-clocked comparator.