# Patent

## Title
PWM comparator and class D amplifier

## Abstract

The present disclosure generally relates to a PWM comparator and a class D amplifier. The PWM comparator described above introduces current feedback mechanism, basing the waveform state of received high frequency triangle signal and the level state of output signal of the PWM comparator, the hysteresis is changing dynamically. In the same resolution, the noise resistance ability of the PWM comparator described above is much better than that of the conventional PWM comparators which has a fixed hysteresis, thus the PWM comparator can work stably even if the duty cycle of output signal is nearly 100%.

## Background
TECHNICAL BACKGROUND
1. Technical Field
The present disclosure relates generally to audio power amplifiers, and more particularly to class D amplifiers.
2. Description of the Related Art
Class D audio power amplifiers are developing rapidly in various applications such as MP3, mobile phone, home theater and so on. They have become the preferred solution in audio systems.
Output power transistors of class D amplifiers work in switch states, their theoretical efficiencies can reach 100%, and their actual efficiencies may be over 90%, compared to actual efficiencies of conventional amplifiers are about 60% or less. The improvement of efficiency may take two advantages: low power consumption and low heat generation. Low heat generation can greatly decrease the size of the heat sink and the area of the PCB of a chip having the class D amplifier. Therefore, the class D amplifier becomes more and more attractive to portable electronics and consumer electronics.
The common modulation principle for a modern class D audio power amplifier is the Pulse Width Modulation (PWM). FIG. 1 is a schematic diagram of a half-bridge class D amplifier incorporating the pulse width modulation. Referring to FIG. 1, the half-bridge class D amplifier includes a pre-operational amplifier, a PWM comparator, a logic circuit, an output power transistor including two MOSFETs, and an external low-pass filter (formed by inductor L and capacitor C) used for restoring amplified audio signal.
The process of the pulse width modulation includes: an audio input signal Vi is amplified by the pre-operational amplifier then to be output as an amplified audio signal Vsin; the amplified audio signal Vsin, is modulated to a pulse signal Vp via a high frequency triangle signal Vosc of the PWM comparator, the pulse signal Vp is as a PWM signal; the PWM signal Vp passes through the logic circuit and drives the output power transistor to output a switch signal Vm; and the switch signal Vm is restored to an audio signal Vout by the LC low-pass filter.
An ideal input-output waveform of the PWM comparator is shown in FIG. 2. According to FIG. 2, when Vsin is higher than Vosc, the PWM signal Vp is output at a high level; when Vsin is lower than Vosc, Vp is output at a low level. This modulation mode is generally called “natural sampling”, in which a frequency fosc of the high frequency triangle signal Vosc is designated as a sampling frequency. In practice, for a balance of efficiency and cost, fosc may be set at about 250 kHz. If fosc is lower than 250 kHz, the efficiency of the class D amplifier may be higher, but a much larger external filter is needed. While fosc is higher than 250 kHz, the situation may be the reverse. The frequency fosc of the high frequency triangle signal Vosc is much higher than that of the audio signal Vsin (20 Hz ̃20 kHz), the pulse width (duty cycle) of the PWM signal Vp in each cycle time of the triangle signal is directly proportional to the amplitude of the amplified audio signal Vsin, which means an audio signal is transformed to pulse signals with different pulse widths.
According to the above description, the PWM comparator is a crucial module in the generation of the PWM signal. According to practical dealt with audio signals, PWM comparators shall have: (1) large bandwidths for achieving high conversion speed; (2) high-gains for achieving high-precision; (3) great noise resistance abilities to ensure that the PWM signal Vp can flip correctly in noisy environment. According to FIG. 2, in a correct modulation process, the PWM signal Vp can only flip from high level to low level on a rising edge of the triangle signal, while the PWM signal Vp can only flip from low level to high level on a falling edge of the triangle signal.
According to FIG. 3, a conventional PWM comparator includes: a pre-amplify circuit, a judge circuit, and an output s circuit. Referring to the judge circuit, a gate of an NMOSFET M6 is connected with a drain of an NMOSFET M7, a gate of the NMOSFET M7 is connected with a drain of the NMOSFET M6, such that a positive feedback is obtained, and the gain of the judge circuit is increased. Further, the connection may achieve to suppress noises by producing a hysteresis. Given that a transconductance gm1 of an NMOSFET M1 for receiving Vsin and a transconductance gm2 of an NMOSFET M2 for receiving Vosc are respectively equal to gm, a width to length ratio (W/L)6,7 of the NMOSFETs M6 and M7 is βB, a width to length ratio of NMOSFETs M5 to M8 (W/L)5,8 is βA, and βB≧βA. In steady states, NMOSFETs M6 and M7 work only in cutoff region or linear region. Given that a drain current ID3 of a PMOSFET M3 is equal to Iss, and a drain current ID4 of a PMOSFET M4 is equal to 0, NMOSFETs M5 and M7 would be active, and NMOSFETs M6 and M8 would be cutoff. With a gradual increase of ID4, ID3 would gradually decrease, and an electric potential of the NMOSFET M7 would increase until the NMOSFET M7 goes into saturation region. At the point that equation (1) below is established:




  
    
      
        
          I
          
            D
            ⁢
            
              
            
            ⁢
            4
          
        
        =
        
          
            
              
                β
                B
              
              2
            
            ⁢
            
              
                (
                
                  
                    V
                    A
                  
                  -
                  
                    V
                    THN
                  
                
                )
              
              2
            
          
          =
          
            
              
                β
                B
              
              
                β
                A
              
            
            ⁢
            
              I
              
                D
                ⁢
                
                  
                
                ⁢
                3
              
            
          
        
      
    
    
      
        (
        1
        )
      
    
  




the threshold point of the current ID4 would be the boundary that the PWM signal Vp flips from high level to low level. After the NMOSFET M7 goes into the saturation region, the NMOSFETs M5 and M7 are cutoff, the NMOSFETs M6 and M8 are active. Similarly, under the situation that ID3 increases and ID4 decreases, at the boundary that the equation (2) below is established, the PWM signal Vp would flip from low level to high level:





  
    
      
        
          I
          
            D
            ⁢
            
              
            
            ⁢
            3
          
        
        =
        
          
            
              β
              B
            
            
              β
              A
            
          
          ⁢
          
            I
            
              D
              ⁢
              
                
              
              ⁢
              4
            
          
        
      
    
    
      
        (
        2
        )
      
    
  




According to equation (1) and equation (3) below:





  
    
      
        
          I
          
            D
            ⁢
            
              
            
            ⁢
            3
          
        
        =
        
          
            
              
                
                  g
                  m
                
                2
              
              ⁢
              
                (
                
                  
                    V
                    sin
                  
                  -
                  
                    V
                    osc
                  
                
                )
              
            
            +
            
              
                I
                ss
              
              2
            
          
          =
          
            
              I
              ss
            
            -
            
              I
              
                D
                ⁢
                
                  
                
                ⁢
                4
              
            
          
        
      
    
    
      
        (
        3
        )
      
    
  




it could be concluded that:





  
    
      
        
          V
          SPH
        
        =
        
          
            
              V
              sin
            
            -
            
              V
              osc
            
          
          =
          
            
              
                
                  I
                  ss
                
                
                  g
                  m
                
              
              ⁢
              
                
                  
                    β
                    A
                  
                  -
                  
                    β
                    B
                  
                
                
                  
                    β
                    A
                  
                  +
                  
                    β
                    B
                  
                
              
            
            =
            
              -
              
                V
                SPL
              
            
          
        
      
    
    
      
        (
        4
        )
      
    
  




wherein VSPH is the flipping upper limit of the PWM comparator (flipping from low level to high level), VSPL is the flipping lower limit of the PWM comparator (flipping from high level to low level).

Referring to FIG. 4, a hysteresis transfer characteristic curve of the PWM signal Vp is shown. The abscissa thereof stands for the value of the difference of Vsin minus Vosc, the ordinate thereof expresses the value of Vp. The changes of the PWM signal Vp are:

when Vsin−Vosc>VSPH, Vp=GND→VCC; when Vsin−Vosc<VSPL, Vp=VCC→GND  (5)

VH=VSPH−VSPL  (6)

However, conventional PWM comparators have these disadvantages:
1) According to equations (4) and (5), once the size of the devices (width to length ratio) is determined, the flipping upper limit VSPH and the flipping lower limit VSPL are fixed, which means the hysteresis section VH is fixed. When PWM are used in audio applications, fixed VH would have disadvantages. When the amplitude of Vosc approaches to Vsin, due to the existence of noises, Vp is prone to flip incorrectly if the VH is set too small. It means that a big noise can make Vp to flip from low level to high level on rising edge of the triangle signal.
2) On the falling edge of the triangle signal Vosc, Vp may flip from high level to low level by mistake. This mistake may happen in the case that the duty cycle of a switch signal is nearly 100%. In that time the triangle signal is in the process of changing from the falling edge to the rising edge. Normally, the triangle signal is obtained through charging and discharging of a capacitor, which would be easy to generate a relatively large noise. If the hysteresis section VH is determined to be too large, the resolution of the PWM comparator will decrease significantly, which would cause a relatively large distortion.
SUMMARY
The present disclosure provides a PWM comparator and a class D amplifier in order to enhance the noise resistance ability of devices.
In order to solve the problem described above, the present disclosure provides a PWM comparator including a pre-amplified circuit, a feedback circuit and a judge circuit. The pre-amplified circuit includes two input-terminals respectively for receiving a high frequency triangle signal and an input audio signal, and the pre-amplified circuit outputs the changes of waveform of the high frequency triangle signal and the input audio signal to the feedback circuit and the judge circuit. The feedback circuit is connected to the pre-amplified circuit and the judge circuit, the feedback circuit comprises an internal current source, the feedback circuit adjusts the current from the current source to the input of the judge circuit, according to the waveform changes of the high frequency triangle signal received from the pre-amplified circuit, and a voltage level of PWM signals of the PWM comparator. The judge circuit is connected to the feedback circuit, the judge circuit obtains the changes of the high frequency triangle signal and the input audio signal based on a change of a input current of the judge circuit, and generates a comparison result according to a comparison of the high frequency triangle signal and the input audio signal.
The present disclosure further provides a class D amplifier including the PWM comparator described above.
The PWM comparator and class D described above have these advantages: the PWM comparator introduces current feedback principle, based on the waveform state of receiving high frequency triangle signal and the level state of output signal of the PWM comparator, the hysteresis changes dynamically. With the same resolution, the noise resistance ability of the PWM comparator described above is much better than that of the conventional PWM comparators which has a fixed hysteresis, thus the PWM comparator can work stably even if the duty cycle of output signal is nearly 100%.



BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a schematic diagram of a half-bridge class D amplifier based on the pulse width modulation;
FIG. 2 shows an ideal waveform of a PWM;
FIG. 3 is a circuit diagram of a conventional PWM comparator;
FIG. 4 shows a state transfer characteristic curve of the PWM comparator shown in FIG. 3;
FIG. 5 is a circuit diagram of a PWM comparator in accordance with an embodiment of the present disclosure;
FIG. 6 shows a state transfer characteristic curve of the PWM comparator showed shown in FIG. 5;
FIG. 7 is a circuit diagram of a PWM comparator in accordance with another embodiment of the present disclosure;
FIG. 8 shows a simulation waveform of state transfer characteristic of the PWM comparator shown in FIG. 7;
FIG. 9 shows a simulation waveform of alternating gain of the PWM comparator shown in FIG. 7;
FIG. 10 shows a simulation waveform of transient of the PWM comparator shown in FIG. 7.



DETAILED DESCRIPTION
The detailed description of the present disclosure will be discussed in the following embodiments, which are not intended to limit the scope of the present disclosure, but still can be adapted for other applications. While drawings are illustrated in details, it is appreciated that the quantity of the disclosed components could be greater or less than that disclosed, except those components with expressly restricted amount.
According to the current disclosure, a current feedback principle can be introduced into the conventional fixed hysteresis-section PWM comparator. The waveform state (on rising edge or falling edge) of the high frequency triangle signal Vosc and the level state (high level or low level) of the PWM signal Vp can be reflected by the current feedback. Further, the hysteresis can be determined basing the state of the high frequency triangle signal Vosc and the PWM signal Vp.
According to FIG. 5, a PWM comparator in accordance with an exemplary embodiment of the present disclosure is shown. The PWM comparator includes a pre-amplified circuit 10, a feedback circuit 20, a judge circuit 30 and an output buffer circuit 40. The pre-amplified circuit 10 has two input-terminals, one for receiving a high frequency triangle signal Vosc, the other for receiving an input audio signal Vsin. The buffer circuit 40 outputs a PWM signal Vp according to a comparison between the high frequency triangle signal Vosc and the input audio signal Vsin.
The pre-amplified circuit 10 includes: a first input NMOSFET M1, a second input NMOSFET M2, a first mirror PMOSFET M31, and a second mirror PMOSFET M41. A gate of the first input NMOSFET M1 is coupled to receive the input audio signal Vsin, drains of the first input NMOSFET M1 and the first mirror PMOSFET M31 are connected together, and sources of the first input NMOSFET M1 and the second input NMOSFET M2 are connected together. A gate of the second input NMOSFET M2 is coupled to receive the high frequency triangle signal Vosc, drains of the second input NMOSFET M2 and the second mirror PMOSFET M41 are connected together. A first current source 11 is coupled between a ground GND and sources of the first input NMOSFET M1 and the second input NMOSFET M2. The current source 11 provides an output current Iss. A gate and a drain of the first mirror PMOSFET M31 are connected directly; and a source of the first mirror PMOSFET M31 is connected with a power source Vcc. A gate and a drain of the second mirror PMOSFET M41 are connected directly, and a source of the second mirror PMOSFET M41 is connected with the power source Vcc.
The feedback circuit 20 includes a second current source 21, a third current source 22, a first switch K1, and a second switch K2. The second current source 21 provides an output current I1, the third current source 22 provides an output current I2, provided that I1 and I2 both equal to I. One terminal of the first switch K1 is connected with the second current source 21. According to the waveform state of the high frequency triangle signal Vosc, the other terminal of the first switch K1 is able to be connected with a first contact terminal a coupling to the gate of the second mirror PMOSFET M41, or a second contact terminal b connecting with the gate of the first mirror PMOSFET M31. One terminal of the second switch K2 is connected with the third current source 22. According to the level state of the PWM signal Vp, the other terminal of the second switch K2 is able to be connected with a third contact terminal c coupling to the gate of the second mirror PMOSFET M41, or a fourth contact terminal d connected with the gate of the first mirror PMOSFET M31. The first switch K1 and the second switch K2 are controlled as follows: when the high frequency triangle signal Vosc is on its rising edge, the first switch K1 is connected to the first contact terminal a; when the high frequency triangle signal Vosc is on its falling edge, the first switch K1 is connected to the second contact terminal b; when the PWM signal Vp is low level, the second switch K2 is connected with the third contact terminal c; when the PWM signal Vp is high level, the second switch K2 is connected with the fourth contact terminal d.
The judge circuit 30 includes a third mirror PMOSFET M3, a fourth mirror PMOSFET M4, a first comparing NMOSFET M5, a second comparing NMOSFET M6, a third comparing NMOSFET M7, and a fourth comparing NMOSFET M8. The source of the third mirror PMOSFET M3 is connected with the power source Vcc, gates of the third mirror PMOSFET M3 and the first mirror PMOSFET M31 are connected together, drains of the third mirror PMOSFET M3 and the first comparing NMOSFET M5 are connected together. The third mirror PMOSFET M3 and the first mirror PMOSFET M31 cooperatively constitutes a mirror current source. A source of the fourth mirror PMOSFET M4 is connected with the power source Vcc, gates of the fourth mirror PMOSFET M4 and the second mirror PMOSFET M41 are connected together, drains of the fourth mirror PMOSFET M4 and the fourth comparing NMOSFET M8 are connected together. The fourth mirror PMOSFET M4 and the second mirror PMOSFET M41 cooperatively constitutes a mirror current source. The gate and the drain of the first comparing NMOSFET M5 are interconnected; the source of the first comparing NMOSFET M5 is grounded. The gate of the second comparing NMOSFET M6 connects to the drain of the third comparing NMOSFET M7, the drain of the second comparing NMOSFET M6 connects the gate of the third comparing NMOSFET M7, and sources of the second comparing NMOSFET M6 and the third comparing NMOSFET M7 are grounded. The gate of the third comparing NMOSFET M7 is connected with the drain of the third mirror PMOSFET M3; the drains of the third comparing NMOSFET M7 and the fourth mirror PMOSFET M4 are connected together. The gate and the drain of the fourth comparing NMOSFET M8 are interconnected; the source of the fourth comparing NMOSFET M8 connects to ground.
The output buffer circuit 40 has two input terminals connected to the gate and the drain of the third comparing NMOSFET M7, respectively, and one output terminal is for outputting the PWM signal Vp.
The analysis for the PWM comparator shown in FIG. 5 is as follows:
Hysteresis of the PWM comparator described above will change by the states of the switches K1 and K2, therefore the PWM comparator is a dynamic hysteresis comparator.
In the feedback circuit, regardless of the states of the switches K1 and K2, the equation (7) as below establishes:

ID3+ID4=2I+ISS  (7)

Designating the width to length ratio (W/L)6,7 of the second comparing NMOSFET M6 and the third comparing NMOSFET M7 equals to βC, width to length ratio (W/L)5,8 of the first comparing NMOSFET M5 and the fourth comparing NMOSFET M8 equals to βA. Incorporating the high frequency triangle signal Vosc and the PWM signal Vp, comprehensively analyze the hysteresis situation of dynamic hysteresis comparator in various switch statuses as follows:
1) When the high frequency triangle signal Vosc is on its rising edge and the PWM signal Vp is at a high level, the switch K1 is connected with the first contact terminal a, the switch K2 is connected with the second contact terminal d, and the below equation (8) establishes:




  
    
      
        
          I
          
            D
            ⁢
            
              
            
            ⁢
            3
          
        
        =
        
          
            
              
                g
                m
              
              2
            
            ⁢
            
              (
              
                
                  V
                  sin
                
                -
                
                  V
                  osc
                
              
              )
            
          
          +
          
            
              I
              SS
            
            2
          
          +
          I
        
      
    
    
      
        (
        8
        )
      
    
  




In this state, the PWM signal Vp can only flip from the high level to the low level, such that only a flipping lower limit VSPL0 should be considered. Referring to the analysis of the fixed hysteresis comparator described above, according to equations (1), (2), (7) and (8), and change βB to βC, there is equation (9) establishes:




  
    
      
        
          V
          
            SPL
            ⁢
            
              
            
            ⁢
            0
          
        
        =
        
          
            
              
                I
                SS
              
              +
              
                2
                ⁢
                I
              
            
            
              g
              m
            
          
          ⁢
          
            
              
                β
                C
              
              -
              
                β
                A
              
            
            
              
                β
                C
              
              +
              
                β
                A
              
            
          
        
      
    
    
      
        (
        9
        )
      
    
  




2) When the high frequency triangle signal Vosc is on its rising edge and the PWM signal Vp is at a low level, the switch K1 is connected with the first contact terminal a, and the switch K2 is connected with the third contact terminal c, and equation (10) as below establishes:




  
    
      
        
          I
          
            D
            ⁢
            
              
            
            ⁢
            3
          
        
        =
        
          
            
              
                g
                m
              
              2
            
            ⁢
            
              (
              
                
                  V
                  sin
                
                -
                
                  V
                  osc
                
              
              )
            
          
          +
          
            
              I
              SS
            
            2
          
        
      
    
    
      
        (
        10
        )
      
    
  




In this state, the PWM signal Vp can only flip from low level to high level, such that only a flipping higher limit VSPH1 should be considered, there is equation (11) as below establishes:




  
    
      
        
          V
          
            SPH
            ⁢
            
              
            
            ⁢
            1
          
        
        =
        
          
            
              
                
                  I
                  SS
                
                +
                
                  2
                  ⁢
                  I
                
              
              
                g
                m
              
            
            ⁢
            
              
                
                  β
                  A
                
                -
                
                  β
                  C
                
              
              
                
                  β
                  C
                
                +
                
                  β
                  A
                
              
            
          
          +
          
            
              2
              ⁢
              I
            
            
              g
              m
            
          
        
      
    
    
      
        (
        11
        )
      
    
  




3) When the high frequency triangle signal Vosc is on the falling edge and the PWM signal Vp is at a low level, the switch K1 is connected with the second contact terminal b, the switch K2 is connected with the third contact terminal c, and the below equation (12) establishes:




  
    
      
        
          I
          
            D
            ⁢
            
              
            
            ⁢
            3
          
        
        =
        
          
            
              
                g
                m
              
              2
            
            ⁢
            
              (
              
                
                  V
                  sin
                
                -
                
                  V
                  osc
                
              
              )
            
          
          +
          
            
              I
              ss
            
            2
          
          +
          I
        
      
    
    
      
        (
        12
        )
      
    
  




In this state, the PWM signal Vp can only flip from low level to high level, such that only a flipping higher limit VSPH0 should be considered, there is the below equation (13) establishes:




  
    
      
        
          V
          
            SPH
            ⁢
            
              
            
            ⁢
            0
          
        
        =
        
          
            
              
                
                  I
                  SS
                
                +
                
                  2
                  ⁢
                  I
                
              
              
                g
                m
              
            
            ⁢
            
              
                
                  β
                  A
                
                -
                
                  β
                  C
                
              
              
                
                  β
                  C
                
                +
                
                  β
                  A
                
              
            
          
          =
          
            -
            
              V
              
                SPL
                ⁢
                
                  
                
                ⁢
                0
              
            
          
        
      
    
    
      
        (
        13
        )
      
    
  




4) When the high frequency triangle signal Vosc is on the falling edge and the PWM signal Vp is at a high level, the switch K1 is connected with the second contact terminal b, the switch K2 is connected with the fourth contact terminal d, and the below equation (14) establishes:




  
    
      
        
          I
          
            D
            ⁢
            
              
            
            ⁢
            3
          
        
        =
        
          
            
              
                g
                m
              
              2
            
            ⁢
            
              (
              
                
                  V
                  sin
                
                -
                
                  V
                  osc
                
              
              )
            
          
          +
          
            
              I
              SS
            
            2
          
          +
          
            2
            ⁢
            I
          
        
      
    
    
      
        (
        14
        )
      
    
  




In this state, the PWM signal Vp can only flip from high level to low level, such that only a flipping lower limit VSPL1 should be considered, expressed as the equation (15) below:




  
    
      
        
          V
          
            SPL
            ⁢
            
              
            
            ⁢
            1
          
        
        =
        
          
            
              
                
                  
                    I
                    SS
                  
                  +
                  
                    2
                    ⁢
                    I
                  
                
                
                  g
                  m
                
              
              ⁢
              
                
                  
                    β
                    C
                  
                  -
                  
                    β
                    A
                  
                
                
                  
                    β
                    C
                  
                  +
                  
                    β
                    A
                  
                
              
            
            -
            
              
                2
                ⁢
                I
              
              
                g
                m
              
            
          
          =
          
            -
            
              V
              
                SPH
                ⁢
                
                  
                
                ⁢
                1
              
            
          
        
      
    
    
      
        (
        15
        )
      
    
  




FIG. 6 shows a state transfer characteristic curve of the PWM comparator as shown in FIG. 5. In FIG. 6, the abscissa stands for the difference of Vsin minus Vosc, the ordinate stands for the value of Vp. Referring together to FIG. 5 and FIG. 6, give equations that VH0=VSPH0−VSPL0, and VH1=VSPH1−VSPL1. According to the above description, in the correct processes of pulse width modulation, the PWM signal Vp can only flip in state 1) and 3), so it is incorrect if the PWM signal Vp flips in state 2) and 4). Therefore, it can be clear that VH0 affects the resolution of the PWM comparator, and VH1 determines the noise resistance ability range of the PWM comparator. Adjusting βC to make it equal to βB, for making VH0 being equal to VH further for ensuring the resolution of the PWM comparator as shown in FIG. 5 to be the same as the resolution of the PWM comparator as shown in FIG. 3. According to equations (9), (11), (13), and (15), the below equation (16) establishes:




  
    
      
        
          V
          
            H
            ⁢
            
              
            
            ⁢
            1
          
        
        =
        
          
            
              V
              
                H
                ⁢
                
                  
                
                ⁢
                0
              
            
            +
            
              
                4
                ⁢
                I
              
              
                g
                m
              
            
          
          =
          
            
              V
              H
            
            +
            
              
                4
                ⁢
                I
              
              
                g
                m
              
            
          
        
      
    
    
      
        (
        16
        )
      
    
  




According to the above description, in the dynamic hysteresis comparator, the hysteresis value which affects the resolution (accuracy) is different from the hysteresis value which affects the noise resistance ability range of the PWM comparator. In the current embodiment, the resolution and the noise resistance ability range are separated to be independent features. According to equation (16), if the resolutions are identical, the PWM comparator as shown in FIG. 5 has a larger noise resistance ability range than that of the conventional fixed hysteresis comparator.
According to FIG. 7, another exemplary embodiment of the PWM comparator of the present disclosure includes a pre-amplified circuit 100, a feedback circuit 200, a judge circuit 300, and an output buffer circuit 400. The pre-amplified circuit 100 has two input-terminals, one is for receiving a high frequency triangle signal Vosc, and the other is for receiving an input audio signal Vsin. The buffer circuit 400 outputs a PWM signal Vp according to a comparison between the high frequency triangle signal Vosc and the input audio signal Vsin.
The pre-amplified circuit 100 includes: a first input NPN transistor Q1, a second input NPN transistor Q2, a first mirror PMOSFET M5a, a second mirror PMOSFET M6a, a first mirror NMOSFET M1a, and a second mirror NMOSFET M2a. The base of the first input NPN transistor Q1 is used for receiving the input audio signal Vsin, the collector of the first input NPN transistor Q1 connects with the drain of the first mirror PMOSFET M5a, emitters of the first input NPN transistor Q1 and the second input NPN transistor Q2 are connected together. The base of the second input NPN transistor Q2 is used for receiving a high frequency triangle signal Vosc, the collector of the second input NPN transistor Q2 is connected with the drain of the second mirror PMOSFET M6a. The first mirror NMOSFET M1a and the second mirror NMOSFET M2a cooperatively constitutes a mirror current source, the drain of the first mirror NMOSFET M1a is connected with one terminal of a current source 101. Both the emitters of the first input NPN transistor Q1 and the second input NPN transistor Q2 are connected with the drain of the second mirror NMOSFET M2a. Sources of the first mirror NMOSFET M1a and the second mirror NMOSFET M2a are grounded.
The other terminal of the current source 101 is connected with a power source Vcc, the current source 101 provides a reference current Ir. The gate and the drain of the first mirror PMOSFET M5a is interconnected. The source of the first mirror PMOSFET M5a is connected with the power source Vcc. The gate and the drain of the second mirror PMOSFET M6a are interconnected, the source of the second mirror PMOSFET M6a is connected with the power source Vcc.
The feedback circuit 200 includes a first feedback NMOSFET M17, a second feedback NMOSFET M18, a third feedback NMOSFET M19, a fourth feedback NMOSFET M20, a third mirror NMOSFET M3a, and a fourth mirror NMOSFET M4a. The gate of the first feedback NMOSFET M17 is used for receiving an external control signal Vctl, the drain of the first feedback NMOSFET M17 is connected with the gate of the second mirror PMOSFET M6a. Both the sources of the first feedback NMOSFET M17 and the second feedback NMOSFET M18 are connected with the drain of the third mirror NMOSFET M3a. The gate of the second feedback NMOSFET M18 is used for receiving a reference voltage Vr, and the drain of the second feedback NMOSFET M18 is connected with the gate of first mirror PMOSFET M5a. The gate of the third feedback NMOSFET M19 is used for receiving the reference voltage Vr, the drain of the third feedback NMOSFET M19 is connected with the gate of the second mirror PMOSFET M6a, and the source of the third feedback NMOSFET M19 is connected with both the source of the fourth feedback NMOSFET M20 and the drain of the fourth mirror NMOSFET M4a. The gate of the fourth feedback NMOSFET M20 is connected with the output-terminal of the output buffer circuit 400 in order to receive the PWM signal Vp, and the drain of the fourth feedback NMOSFET M20 is connected with the gate of the first mirror PMOSFET M5a. The third mirror NMOSFET M3a and the first mirror NMOSFET M1a cooperatively constitutes a mirror current source. The fourth mirror NMOSFET M4a and the first mirror NMOSFET M1a cooperatively constitutes a mirror current source. The sources of the third mirror NMOSFET M3a and the fourth mirror NMOSFET M4a are connected to ground.
The judge circuit 300 includes a third mirror PMOSFET M7a, a fourth mirror PMOSFET M8a, a first comparing NMOSFET M9, a second comparing NMOSFET M10, a third comparing NMOSFET M11, a fourth comparing NMOSFET M12, a fifth mirror NMOSFET M13, a sixth mirror NMOSFET M14, a fifth mirror PMOSFET M15, and a sixth mirror PMOSFET M16. The source of the third mirror PMOSFET M7a is connected with the power source Vcc, the gate of the third mirror PMOSFET M7a is connected with the gate of the first mirror PMOSFET M5a, and the drains of the third mirror PMOSFET M7a and the first comparing NMOSFET M9 are connected together. The third mirror PMOSFET M7a and the first mirror PMOSFET M5a cooperatively constitutes a mirror current source. The source of the fourth mirror PMOSFET M8a is connected with the power source Vcc, the gates of the fourth mirror PMOSFET M8a and the second mirror PMOSFET M6a are connected together, and the drains of the fourth mirror PMOSFET M8a and the second comparing NMOSFET M10 are connected together. The fourth mirror PMOSFET M8a and the second mirror PMOSFET M6a cooperatively constitutes a mirror current source. The gate and the drain of the first comparing NMOSFET M9 are interconnected; the source of the first comparing NMOSFET M9 connects to ground. The sources of the third comparing NMOSFET M11 and the fourth comparing NMOSFET M12 are connected to ground; the gate of the third comparing NMOSFET M11 connects with the drain of the fourth comparing NMOSFET M12, and the gate of the fourth comparing NMOSFET M12 connects with the drain of the third comparing NMOSFET M11. The gate of the fourth comparing NMOSFET M12 is connected with the drain of the third mirror PMOSFET M7a, and the drains of the fourth comparing NMOSFET M12 and the fourth mirror PMOSFET M8a are connected together. The gate and the drain of the second comparing NMOSFET M10 are interconnected, and the source of the second comparing NMOSFET M10 connects to ground. The fifth mirror NMOSFET M13 and the first comparing NMOSFET M9 cooperatively constitutes a mirror current source, the source of the fifth mirror NMOSFET M13 connects to ground, and the drains of the fifth mirror NMOSFET M13 and the sixth mirror PMOSFET M16 are connected together. The sixth mirror NMOSFET M14 and the second comparing NMOSFET M10 cooperatively constitutes a mirror current source, the source of the sixth mirror NMOSFET M14 connects to ground, and the drains of the sixth mirror NMOSFET M14 and the fifth mirror PMOSFET M15 are connected together. The fifth mirror PMOSFET M15 and the sixth mirror PMOSFET M16 cooperatively constitutes a mirror current source, and the sources of the fifth mirror PMOSFET M15 and the sixth mirror PMOSFET M16 connect with power source Vcc.
The output buffer circuit 400 includes: a first inverter 403, a Schmitt trigger 401, and a second inverter 402 that are connected in series. The input-terminal of the first inverter 403 is connected with the drain of the sixth mirror PMOSFET M16, and the second inverter 402 is coupled for inputting the PWM signal Vp.
The PWM comparator showed in FIG. 7 is described as follows:
In the pre-amplified circuit 100, the first input NPN transistor Q1 and the second input NPN transistor have a lower imbalance voltage and a higher gain comparing with the MOS devices which have the same layout area.
The pre-amplified circuit 100 and the judge circuit 300 have a CMOS operational transconductance amplifier (OTA) structure, the interconnection node are all low-resistance nodes except for the drains of the sixth mirror PMOSFET M16 and the fifth mirror NMOSFET M13, thus implementing a large bandwidth.
The output buffer circuit 400, formed by two inverters and a Schmitt trigger, functions as follows: (1) shaping the output signal of the judge circuit 300 to reduce the conversion time of the output pulse signal (flipping from high to low or from low to high); (2) increasing driving capability; (3) isolating external load; (4) the Schmitt trigger has a relatively good noise resistance ability.
The fourth feedback NMOSFETs M17 ̃M20 in the feedback circuit 200 function similar to the first switch K1 and the second switch K2, and the waveform state of the external control signal Vctl corresponds with that of the high frequency triangle signal Vosc. While the high frequency triangle signal Vosc is in rising edge, the external control signal Vctl is high level; while the high frequency triangle signal Vas, is on the falling edge, the external control signal Vctl is low level.
In order to verify the performance of the PWM comparator as shown in FIG. 7, the size of the devices are designed as follows: the width to length ratios of the first mirror NMOSFET M1a, the third mirror NMOSFET M1a, and the fourth mirror NMOSFET M4a are identical; the width to length ratio of the second mirror NMOSFET M2a is four times as that of the first mirror NMOSFET M1a; the width to length ratios of the first mirror PMOSFET M5a, the second mirror PMOSFET M6a, the third mirror PMOSFET M7a, and the fourth mirror PMOSFET M8a are identical, which are greater than that of the second mirror NMOSFET M2a; the width to length ratios of the first comparing NMOSFET M9, the second comparing NMOSFET M10, the fifth mirror NMOSFET M13, and the sixth mirror NMOSFET M14 are identical, which are greater than that of the first mirror NMOSFET M1a and less than that of the second mirror NMOSFET M2a; the width to length ratios of the third comparing NMOSFET M11 and the fourth comparing NMOSFET M12 are identical, which are greater than that of the first comparing NMOSFET M9 and less than that of the second mirror NMOSFET M2a; the width to length ratios of the fifth mirror PMOSFET M15 and the sixth mirror PMOSFET M16 are identical, which are greater than that of the first mirror PMOSFET M5a; the width to length ratios of the first feedback NMOSFET M17 to the fourth feedback NMOSFET M20 are identical, which are less than that of the first mirror NMOSFET M1a. And further, a suitable reference current and reference voltage is set.
Basing the size design and the reference current and reference voltage settings described above, FIG. 8 shows a simulation waveform of state transfer characteristic of the PMW comparator simulated in Spectre. The input in the Vosc terminal is a 1 ̃4V triangle signal, and the input in the Vsin terminal is an analog audio signal. According to FIG. 8, when the PWM comparator is in state 1) or state 3), VSPH0 is about 2.51V and VSPL0 is about 2.49V, and therefore VH0 equals about 20 mV; when the PWM comparator is in state 2) or state 4) (incorrect flipping), VSPH1 is about 2.54V and VSPL1 is about 2.46V, therefore VH1 equals about 80 mV. It can be seen that, with identical resolutions (VH0 equals VH), compared with the PWM comparator as shown in FIG. 3, the comparator as shown in FIG. 7 has a larger 60 mV noise resistance ability range, therefore the capability of the PWM comparator has been enhanced greatly.
FIG. 9 shows a simulation waveform of the alternating gain of the PWM comparator as shown in FIG. 7. According to FIG. 9, the gain of the PWM comparator attains to 79 dB, the 3 dB bandwidth is larger than 1 MHz, and the unity gain bandwidth is larger than 10 MHz, which can meet the design requirements of high-speed and high-precision.
FIG. 10 shows a simulation waveform of transient of the PWM comparator of FIG. 7. According to FIG. 10, a sinusoidal signal, with a common-mode voltage of 2.5V, an amplitude of 0.5V, and a frequency of 10 kHz, is applied at the Vsin terminal, its; and a triangle signal, with a frequency of 250 kHz, and an amplitude of 2 ̃3V, is applied at the Vosc terminal. During the whole modulation process, the PWM comparator of FIG. 7 works stably even if the duty cycle is nearly 100%.
In addition, the present disclosure further provides a class D amplifier incorporating the PWM comparator as described above, the pre-circuit and after-circuit of the PWM comparator in the class D amplifier can be conventional, which will not be described in detail here.
In summary, by introducing two ways of current feedback, the PWM comparator of the present disclosure has a dynamic hysteresis. Under identical resolutions, the noise resistance ability of the present disclosure is much better than conventional fixed-hysteresis comparator, thus the PWM comparator can work stably even if the duty cycle of the output signal is nearly 100%.
Although the invention has been described in language specific to structural features and/or methodological acts, it is to be understood that the invention defined in the appended claims is not necessarily limited to the specific features or acts described. Rather, the specific features and acts are disclosed as example forms of implementing the claimed invention.

## Claims

1. A PWM comparator comprising a preamplifier circuit, a feedback circuit and a judge circuit; wherein
the preamplifier circuit comprises two input-terminals respectively for receiving a high frequency triangle signal and an input audio signal, and the preamplifier circuit outputs the changes of waveform of the high frequency triangle signal and the input audio signal to the feedback circuit and the judge circuit;
the feedback circuit is connected to the preamplifier circuit and the judge circuit, the feedback circuit comprises an internal current source, the feedback circuit adjusts the current from the current source to the input of the judge circuit, according to the waveform changes of the high frequency triangle signal received from the pre amplified preamplifier circuit, and a voltage level of PWM signals of the PWM comparator;
the judge circuit is connected to the feedback circuit, the judge circuit obtains the changes of the high frequency triangle signal and the input audio signal based on a change of a input current of the judge circuit, and generates a comparison result according to a comparison of the high frequency triangle signal and the input audio signal.

2. The PWM comparator according to claim 1, further comprises an output buffer circuit connected with the feedback circuit and the judge circuit for producing and outputting a PWM signal according to the comparison result.

3. The PWM comparator according to claim 2, wherein the preamplifier circuit comprises: a first input NMOSFET, a second input NMOSFET, a first mirror PMOSFET, and a second mirror PMOSFET; wherein
the first input NMOSFET has a gate for receiving the input audio signal, a drain connected to a drain of the first mirror PMOSFET, and a source connected to a source of the second input NMOSFET;
the second input NMOSFET has a gate for receiving the high frequency triangle signal, a drain to be connected to a drain of the second mirror PMOSFET;
a first current source for being connected between ground and sources of the first input NMOSFET and the second input NMOSFET;
the first mirror PMOSFET has a gate and a drain interconnected to each other, and a source connected to a power source;
the second mirror PMOSFET has a gate and a drain interconnected to each other, and a source connected to the power source.

4. The PWM comparator according to claim 3, wherein the feedback circuit comprises: a second current source, a third current source, a first switch, and a second switch; wherein
one terminal of the first switch is connected to the second current source, the other terminal is able to be connected to a first contact terminal or a second contact terminal based on the waveform of the high frequency triangle signal; wherein the first contact terminal is connected to the gate of the second mirror PMOSFET, the second contact terminal is connected to the gate of the first mirror PMOSFET;
one terminal of the second switch is connected to the third current source, the other terminal is able to be connected to a third contact terminal or a fourth contact terminal based on the voltage level of the pulse width signal; wherein the gate of the second mirror PMOSFET is connected to the third contact terminal, the gate of the first mirror PMOSFET is connected to the fourth contact terminal.

5. The PWM comparator according to claim 4, wherein the first switch is switchably connected to the first contact terminal at a rising edge of the high frequency triangle signal, and switchably connected to the second contact terminal at the falling edge of the high frequency triangle signal; the second switch is switchably connected to the third contact terminal at the low level of the PWM signal, and switchably connected to the fourth contact terminal at the high level of the PWM signal.

6. The PWM comparator according to claim 4, wherein the judge circuit comprises: a third mirror PMOSFET, a fourth mirror PMOSFET, a first comparing NMOSFET, a second comparing NMOSFET, a third comparing NMOSFET, and a fourth comparing NMOSFET, wherein
the third mirror PMOSFET has a source connected to the power source, a gate connected to a gate of the first mirror PMOSFET, and a drain connected to a drain of the first comparing NMOSFET;
the fourth mirror PMOSFET has a source connected to the power source, a gate connected to a gate of the second mirror PMOSFET, and a drain connected to a drain of the fourth comparing NMOSFET;
the drain and the gate of the first comparing NMOSFET are interconnected with each other, the source of the first comparing NMOSFET is connected to ground;
the second comparing NMOSFET has a gate connected to the drain of the third comparing NMOSFET, a drain of the second comparing NMOSFET connected to the gate of the third comparing NMOSFET, the sources of the second comparing NMOSFET and the third comparing NMOSFET are connected to ground;
the gate of the third comparing NMOSFET is connected to the drain of the third mirror PMOSFET, the drains of the third comparing NMOSFET and the fourth mirror PMOSFET are interconnected with each other;
the drain and gate of the fourth comparing NMOSFET are interconnected with each other, the source of the fourth comparing NMOSFET is connected to ground.

7. The PWM comparator according to claim 6, wherein the output buffer circuit has two input-terminals connected to the gate and the drain of the third comparing NMOSFET respectively, and one output-terminal for outputting the PWM signal.

8. The PWM comparator according to claim 1, wherein the preamplifier circuit and the judge circuit have a CMOS operational transconductance amplifier structure.

9. The PWM comparator according to claim 2, wherein the preamplifier circuit comprises: a first input NPN transistor, a second input NPN transistor, a first mirror PMOSFET, a second mirror PMOSFET, a first mirror NMOSFET, and a second mirror NMOSFET; wherein
the first input NPN transistor has a base for receiving the input audio signal, a collector connected to a drain of the first mirror PMOSFET, and an emitter connected to an emitter of the second input NPN transistor;
the second input NPN transistor has a base for receiving a high frequency triangle signal and a collector connected to the drain of the second mirror PMOSFET;
the emitters of the first input NPN transistor and the second input NPN transistor are connected to drain of the second mirror NMOSFET, the sources of the first mirror NMOSFET and the second mirror NMOSFET are connected to ground;
the first mirror PMOSFET has a gate and a drain interconnected with each other;
the second mirror PMOSFET has a gate and a drain interconnected with each other.

10. The PWM comparator according to claim 9, wherein the feedback circuit comprises: a first feedback NMOSFET, a second feedback NMOSFET, a third feedback NMOSFET, a fourth feedback NMOSFET, a third mirror NMOSFET, and a fourth mirror NMOSFET; wherein
the first feedback NMOSFET has a gate for receiving a external control signal, a drain connected to the gate of the second mirror PMOSFET, the sources of the first feedback NMOSFET and the second feedback NMOSFET are connected to a drain of the third mirror NMOSFET;
the second feedback NMOSFET has a gate for receiving a reference voltage, a drain connected to the gate of first mirror PMOSFET;
the third feedback NMOSFET has a gate for receiving the reference voltage, a drain connected to the gate of the second mirror PMOSFET, a source connected to the sources of the fourth feedback NMOSFET and drain of the fourth mirror NMOSFET;
the fourth feedback NMOSFET has a gate connected to the output-terminal of the output buffer circuit, a drain connected to the gate of the first mirror PMOSFET;
sources of the third mirror NMOSFET and the fourth mirror NMOSFET are connect to ground.

11. The PWM comparator according to claim 10, wherein the judge circuit comprises: a third mirror PMOSFET, a fourth mirror PMOSFET, a first comparing NMOSFET, a second comparing NMOSFET, a third comparing NMOSFET, a fourth comparing NMOSFET, a fifth mirror NMOSFET, a sixth mirror NMOSFET, a fifth mirror PMOSFET, and a sixth mirror PMOSFET; wherein
the third mirror PMOSFET has a gate connected to the gate of the first mirror PMOSFET, a drain connected to the drain of the first comparing NMOSFET;
the fourth mirror PMOSFET has a gate connected to the gate of the second mirror PMOSFET, a drain connected to a drain of the second comparing NMOSFET;
a gate and a drain of the first comparing NMOSFET are interconnected with each other, a source of the first comparing NMOSFET connects to ground;
sources of the third comparing NMOSFET and the fourth comparing NMOSFET connect to ground, a gate of the third comparing NMOSFET connects with a drain of the fourth comparing NMOSFET, a gate of the fourth comparing NMOSFET is connected to a drain of the third comparing NMOSFET, the gate of the fourth comparing NMOSFET is connected with the drain of the third mirror PMOSFET, the drains of the fourth comparing NMOSFET and the fourth mirror PMOSFET are connected together;
the drain and gate of the second comparing NMOSFET are interconnected with each other, source of the second comparing NMOSFET is connected to ground;
a source of the fifth mirror NMOSFET is connected to ground, drains of the fifth mirror NMOSFET and the sixth mirror PMOSFET are interconnected with each other;
a source of the sixth mirror NMOSFET is connected to ground, drains of the sixth mirror NMOSFET and the fifth mirror PMOSFET are interconnected with each other.

12. The PWM comparator according to claim 11, wherein the output buffer circuit comprises: a first inverter, a Schmitt trigger, and a second inverter connected in series successively; an input-terminal of the first inverter is connected to the drain of the sixth mirror PMOSFET; the second inverter outputs the PWM signal.

13. The PWM comparator according to claim 11, wherein the first mirror NMOSFET, the third mirror NMOSFET, and the fourth mirror NMOSFET have the same width to length ratio; the width to length ratio of the second mirror NMOSFET is four times as that of the first mirror NMOSFET; The first mirror PMOSFET, the second mirror PMOSFET, the third mirror PMOSFET, and the fourth mirror PMOSFET have the same width to length ratio, which is greater than that of the second mirror NMOSFET; the first comparing NMOSFET, the second comparing NMOSFET, the fifth mirror NMOSFET, and the sixth mirror NMOSFET M14 have the same width to length ratio, which is greater than that of the first mirror NMOSFET and less than that of the second mirror NMOSFET; the third comparing NMOSFET and the fourth comparing NMOSFET have the same width to length ratio, which is greater than that of the first comparing NMOSFET and less than that of the second mirror NMOSFET; the fifth mirror PMOSFET has the same width to length ratio as that of the sixth mirror PMOSFET, which is greater than that of the first mirror PMOSFET; the first feedback NMOSFET to the fourth feedback NMOSFET have the same width to length ratio, which is less than that of the first mirror NMOSFET.

14. A class D amplifier comprising the PWM comparator according to claim 1.