A national laboratory of the U.S. Department of Energy 

Office of Energy Efficiency & Renewable Energy 

National Renewable Energy Laboratory 

Innovation for Our Energy Future 

Design and Test of a Variable Subcontract Report 

NREL/SR-500-39715 Speed Wind Turbine System July 2006 

Employing a Direct Drive Axial 

Flux Synchronous Generator 

October 29, 2002 — December 31, 2005 

T.A. Lipo and P. Tenca 

University of Wisconsin 

Madison, Wisconsin

NREL is operated by Midwest Research Institute ● Battelle Contract No. DE-AC36-99-GO10337 
Design and Test of a Variable 

Speed Wind Turbine System 

Employing a Direct Drive Axial 

Flux Synchronous Generator 

October 29, 2002 — December 31, 2005 

T.A. Lipo and P. Tenca 

University of Wisconsin 

Madison, Wisconsin 

NREL Technical Monitor: A. Laxson 

Prepared under Subcontract No. XCX-3-32227-04 

Subcontract Report 

NREL/SR-500-39715 

July 2006 

National Renewable Energy Laboratory

1617 Cole Boulevard, Golden, Colorado 80401-3393 

303-275-3000 • www.nrel.gov 

Operated for the U.S. Department of Energy 

Office of Energy Efficiency and Renewable Energy 

by Midwest Research Institute • Battelle 

Contract No. DE-AC36-99-GO10337
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i 

Table of Contents 

Subject of the Research Project ................................................................................................................... 1

Control Principle............................................................................................................................................ 4

Controller of the CSI Bridges and DC-Link Current ................................................................................. 4

Control of the Active Harmonic Compensator.......................................................................................... 8

Structure and Primary Features of the PM Axial Flux Generator ............................................................... 14

Overview and Features of the Chosen DSP-Based Control System.......................................................... 16

Prototype and Experimental Results........................................................................................................... 17

Suggestions for Future Work ...................................................................................................................... 40

Conclusions................................................................................................................................................. 40

Published Papers Related to the Project Activities..................................................................................... 40

Appendix ..................................................................................................................................................... 41

List of Figures 

Figure 1. Current source topology for wind turbines highlighting the structure of the interconnect with 

the grid.......................................................................................................................................2

Figure 2. Alternative connection utilizing intergroup reactor.....................................................................3

Figure 3. Illustrating angles φ1, φ2 , α, and β............................................................................................5

Figure 4. Amplitudes of the 5th and 7th harmonics versus the phase-control angle ...............................6

Figure 5. Simulated converter waveforms (bottom) for α = 167°. The waveforms in the upper plots are 

bridge 1 DC-link voltage (blue), bridge 2 DC-link voltage (red), Vdc (magenta), and average 

total DC-link voltage <Vdc> (green) ..........................................................................................6

Figure 6. Operational principle characterizing the controller of the main conversion chain for 

β = 0...........................................................................................................................................8

Figure 7. Operational principle characterizing the controller for the active harmonic compensator.........9

Figure 8. Grid line currents (blue) corrected by the VSI .........................................................................10

Figure 9. Spectrum of the corrected grid line currents ...........................................................................11

Figure 10. Voltages influencing the DC-link current .................................................................................11

Figure 11. Regulation performed by the DC-link controller ......................................................................12

Figure 12. Instantaneous power delivered to the grid ..............................................................................12

Figure 13. Simulated generator phase voltages.......................................................................................13

Figure 14. Reference and actual grid line currents...................................................................................13

Figure 15. Reference and actual VSI line currents...................................................................................14

Figure 16. Mechanical drawings of the PM generator frame....................................................................15

Figure 17. Waveforms of two line-to-line back-EMFs...............................................................................15

Figure 18. Self-contained frame hosting the XC 2000 system .................................................................16

Figure 19. XCS 2000 controller board ......................................................................................................16

Figure 20. XCI 2000 four-phase converter control card ...........................................................................17

Figure 21. XCI 2020 universal input/output card ......................................................................................17

Figure 22. Diagonal view of the experimental setup with the Variac and drive frame in the foreground .18

Figure 23. Rear view showing the two CSI bridges, DSP controller, and some of the voltage and current 

probes connected to the instrumentation ................................................................................19

Figure 24. Diagonal view of the setup showing the dynamometer and the PM axial flux 

generator .................................................................................................................................20

Figure 25. Detail of the control panel for the entire prototype. The PC is used to program and 

set the references to the DSP controller, as well as to collect the measurement from the 

oscilloscopes ...........................................................................................................................21

Figure 26. The interconnection transformer, axial flux PM generator, and dynamometer (prime mover 

emulating the propeller of the wind turbine) ............................................................................22

Figure 27. Drive for the dynamometer......................................................................................................23

Figure 28. PM axial flux generator (left) coupled with the dynamometer (right).......................................24

Figure 29. The PM axial flux generator built by Light Engineering...........................................................25

Figure 30. Diode bridge (bottom center) together with the contactors and fuses for DC-link and grid ....26
ii 

Figure 31. Protection circuit for the DC-link inductor described in [3] and reported in the Appendix.......27

Figure 32. P. Tenca machining parts on the center lathe.........................................................................27

Figure 33. P. Tenca machining parts on the milling machine...................................................................28

Figure 34a. α = 167°, β = 0°, Vll = 122VRMS, <Vdc> = 325 V, Idc = 15 A...............................................29

Figure 34b. α = 167°, β = 40° leading, Vll = 120VRMS, <Vdc> = 248 V, Idc = 15 A................................29

Figure 34c. α = 167°, β = 40° lagging, Vll = 120VRMS, <Vdc> = 248V, Idc = 15 A.................................30

Figure 34d. α = 169°, β = 89.9° leading, Vll = 80.8VRMS, <Vdc> = 10.4 V, Idc = 10 A (10 A/div.) .........31

Figure 35. α = 167.8°. Zero of 7th harmonic.............................................................................................32

Figure 36. α = 167°. 5th and 7th harmonics minimized...........................................................................33

Figure 37. α = 164.4°. Zero of 5th harmonic.............................................................................................33

Figure 38. α = 143.9°. Zero of 7th harmonic.............................................................................................34

Figure 39. α = 129.5°. Zero of 5th harmonic.............................................................................................34

Figure 40. α = 120°. Zero of 7th harmonic................................................................................................35

Figure 41. α = 111°. 5th harmonic greater than the 7th ...........................................................................35

Figure 42. α = 94.7°. The grid phase current is negligible........................................................................36

Figure 43. β = 30°. PF leading..................................................................................................................36

Figure 44. β = -30°. PF lagging.................................................................................................................37

Figure 45. β = 60°. PF leading..................................................................................................................37

Figure 46. β = -64.6°. PF lagging..............................................................................................................38

Figure 47. β = 89.9°. PF leading..............................................................................................................38

Figure 48. Transient from 1 A to 10 A of DC-link current at <Vdc> = 328 V, 1464 rpm...........................39

Figure 49. Transient from 10 A to 1 A of DC-link current at <Vdc> = 328 V, 1464 rpm...........................39

List of Tables 

Table 1. Values of the Phase-Control Angle φ Reducing the Harmonic Content of the Grid Line 

Current.......................................................................................................................................5
1 

Subject of the Research Project 

The goal of this funded research project is the definition, analytical investigation, modeling, and 

prototype realization of a current-source conversion topology tailored to high-power wind 

turbines. The important features that such a topology should possess for a wind turbine 

application can be summarized by the following points: 

1. The converter must be capable of delivering power grid line current whose fundamental 

power factor, defined with respect to the corresponding phase voltages, is capable of being 

controlled. It is usually desirable to control the power factor at unity, but on occasion it may 

be desirable to generate reactive power of either polarity. 

2. The harmonic content of the grid line current must be reduced to a reasonable value. (The 

desired goal is defined by the IEEE 519-1992 standards.) 

3. The efficiency of the overall conversion chain—including the generator—is very important 

because it is desirable to extract as much energy as possible from the “free” wind. 

4. The wind turbine generator usually costs at least several times more than the converter, 

which, in turn, represents about 10% of the wind turbine total cost. This means that a 

significant increase in converter cost can be justified if the proposed topology can 

significantly reduce the generator cost. 

5. Failures in the conversion chain contribute to about one third of the total turbine maintenance 

costs during its lifetime. Consequently it is important to use a reliable topology that is highly 

fault-tolerant and redundant in case of component failure. 

6. The points from 3 to 5 above are even more significant for high-power turbines, while point 

2 can eventually be relaxed at such power levels because several turbines present in a large 

wind farm can be interconnected using appropriate transformers in a manner capable of 

reducing the harmonics at the interconnection point of the wind farm and the grid. 

All these considerations have been evaluated together and have led the investigators to define the 

conversion concept illustrated in Figure 1, where the square boxes identify topological structures 

organized into common three-phase bridges. The topology is based on a current-source concept 

whose intrinsic nature exploits the distance between the generator and the power grid 

characterizing the wind turbines to realize either a portion or even all of the necessary DC-link 

inductance. Only two cables are needed between the nacelle and ground, and only the two most 

reliable modules (diode bridge and generator) are located in the nacelle, thereby reducing the 

weight of the nacelle and the overall maintenance costs. The average power is converted entirely 

by the two series-connected, current-source inverters 1 and 2 employing fully controllable 

switches with bidirectional voltage blocking capability. Nevertheless, it must be highlighted that 

one of the bridges can use thyristors in several applications where a reduced range (including 

unity) for the grid fundamental power factor is sufficient, thereby reducing the overall cost of the 

system. 
2 

Figure 1. Current source topology for wind turbines highlighting the structure of the interconnect 

with the grid

The topology shown in Figure 1 is especially tailored for high-power wind turbines, but the 

inverter-side solution, used for the interconnection with the grid, can be employed in various 

other high-power applications. Inverters 1 and 2 have the usual topological structure of a threephase bridge, and they are switched at line frequency, explicitly avoiding pulse-width 

modulation (PWM), thereby reducing switching losses and allowing the use of high-power but 

relatively slow devices such as gate turn off thyristors (GTOs) or integrated gate-commutated 

thyristors (IGCTs). As consequence of this choice, the efficiency of the main conversion chain 

(Generator – diode bridge – two current-source converters in series) is improved. A significant 

part of the project has been focused on the power quality aspects and flexibility in reactive power 

generation by controlling inverters 1 and 2 in a particular manner. The grid currents ia, ib, and ic

originate from the inverter line currents following the algebraic relations, which are consequence 

of the Y-Y-∆ structure of the interconnection transformer to the high voltage utility grid. 

The control methodology adopted for the main conversion chain can deliver unity power factor 

at the grid (transformer primary) under any load condition. Additionally, the operating point of 

the bridges and the generator design can be matched to minimize the 5th and 7th harmonics of the 

grid currents when the turbine operates at nominal power. The undesired harmonics are further 

reduced by the active harmonic compensator, shown in green in Figure 1, which does not deliver 

any active power. Its control technique is based on the real-time minimization of the “distance” 

(as defined in functional analysis) between the desired reference and actual line currents [1]. The 

use of low-cost standard thyristors is possible in either or both bridges provided that well-known 

commutation circuits are used (for example, ASCI – auto sequentially commutated inverter). 

This renders the topology suitable for very-high-power wind turbines and potentially improves 

its reliability due to the well known ruggedness characterizing the thyristors. It is important to 

observe that an eventual failure of the harmonic compensator (which constitutes the most critical 

sub-system from the reliability point of view) does not prevent, a-priori, the system from 

continuing to operate. Indeed, the harmonic compensator can be completely disconnected and 

the topology would remain capable of delivering the full rated power via the two current-source 

bridges accepting a reduction in the power quality. Even a failure occurring in one of the two 

current source bridges could permit a reduced power operation by short-circuiting the damaged 
3 

bridge and adjusting the blade pitch of the turbine to obtain the suitable generator voltage for 

this uncommon condition. If the harmonic compensator remained operational during this fault in 

the main conversion chain, it could still be used to improve the harmonic content and, in 

principle, even supply some reactive power if connected properly to the transformer secondary 

of the current-source bridge that is still operating. 

The topology shown in Figure 1 can also be reconfigured to test other proposed solutions. For 

example, the “indirect synthesis” topology [2] can be obtained by shorting and keeping one 

bridge inactive (the bridge using thyristors, if this were the case) while the static harmonic 

compensator is substituted with a Y-connected inductor-capacitor filter (L-C filter). If it is 

desirable, one can also connect the two current-source bridges in parallel at the direct current 

(DC) link by providing a three-terminal intergroup reactor instead of a simple inductor. This 

solution would leave unaffected the behavior on the alternating current (AC) side of the 

conversion chain while requiring half of the phase generator voltage and the double its line 

current (when compared to the series-connection solution), as shown in Figure 2. In conclusion, 

the prototype of the aforementioned chosen circuit realizes not only one single specific 

topological solution but also constitutes a reconfigurable test bench that allows exploration of 

different topologies for wind turbines. 

Figure 2. Alternative connection utilizing intergroup reactor

The choice of the tools used to model the selected topology fell on the combination of 

Matlab/Simulink (by Mathworks) and Simplorer (by Ansoft), mainly because Simplorer 

possesses an interface which provides the strategic ability to simulate in parallel and 

communicate with Matlab/Simulink. In this manner, it is possible to devote the simulation of the 

topology to a true circuit simulator (Simplorer) while keeping the ability to implement its control 

system entirely in Simulink. Furthermore, all desired simulation results can be transferred to 

Matlab for their subsequent evaluation, post-processing, and storage. 
4 

Control Principle

This section gives an overview of the operational principles at the basis of the overall controller 

for the topology shown in Figure 1. It is possible to identify two main control tasks in the 

systems which are voluntarily separate from each other for the sake of the fault-tolerance of the 

converter. This also avoids unnecessary mutual dependencies among control loops which 

usually lead to behavior that is difficult to diagnose and correct. 

Controller of the CSI Bridges and DC-Link Current 

The first and most important control task is the regulation of the phase angle of the two current 

source inverter (CSI) bridges 1 and 2, and the consequent regulation of the DC-link current 

around the desired constant value associated with the actual speed of the turbine via its planned 

“power vs. rotational speed” curve. 

The inverters 1 and 2 are operated independently using the conventional phase-control technique, 

which leads to the displacement of the fundamental line currents with respect to the related 

fundamental phase voltages by the controlled angles φ1 and φ2 (for the bridges 1 and 2 

respectively). With the use of fully controllable devices, the control angles can range over the 

interval [-π, π], thereby allowing not only lagging but leading fundamental power factor 

operation. This is a key distinctive feature of this solution versus the case in which only 

thyristors are used in one of the two bridges. Under the assumptions of instantaneous current 

commutation between any two switches and an ideal transformer, Equations (1) and (2) below 

constitute the general expressions for the modulus and absolute phase, respectively, of the 

complex Fourier coefficient Cn of the grid line current ia. In Equations (1) and (2), α = (φ1 - φ2)/2, 

β = (φ1 + φ2)/2 , n is the harmonic order, and Idc is DC-link current (assumed constant). 

From these equations, it may be seen that the amplitudes of the harmonics depend only on α; i.e., 

the difference between φ1 and φ2, while the absolute phase depends only on β, i.e., the sum of φ1

and φ2. From Equation (2) it can inferred that β performs a pure phase displacement of the grid 

line current waveform because the absolute phases of all harmonics are linearly dependent on β

through the harmonic order n. Figure 3 shows the aforementioned angles together with the 

phasors of the fundamental of the currents in the corresponding phases of the bridges 1 (blue) 

and 2 (red). 

 ( ( ) ) ( ) 2 4 1 1 2 2cos cos cos

3 6

n

n dc

n n C I n

n

π π

α

π

⎛ ⎛ ⎞ ⎞ = ⋅ ⋅ −− ⋅ − ⋅ ⋅ ⎜ ⎟ ⎜⎟ ⎝ ⎝ ⎠ ⎠ (1)

 ( ) 1 cos(4 / 3) 1 arg tan

sin(4 /3) n

n C n

n

π β

π

− ⎛ − ⎞

= + ⎜ ⎟ ⎝ ⎠ (2) 

 6

cos cos Average value of llpk

dc

V

V V α β dc

π

⋅ = ⋅⋅ (3) 

Vllpk is the amplitude of the line-to-line voltage at the transformer having identical secondaries. 
5 

φ1

φ2

α

-α β

α = φ1 - φ2

2

β = φ1 + φ2

2

Figure 3. Illustrating angles φ1, φ2 , α, and β

From the previous observations, it follows that the angle α can be used to control the amplitudes 

of the harmonic components present in the grid line currents while the angle β can be used to 

independently control the fundamental power factor of these currents with respect to the related 

grid phase voltages. In [1], it was demonstrated that the amplitude of the 5th or 7th harmonic of 

the line currents, computable via Equation (1), can be nullified for specific values of the angle α

while always maintaining an essentially unity fundamental grid power factor. Furthermore, it 

was shown in [1] that by selecting α = 0.914·π (164.52°), one obtains a minimum sum of the 

average power associated with the 5th and 7th harmonics, highly desirable for power quality. 

These results are summarized in Table 1 and Figure 4. It follows that by controlling α to remain 

close to 0.914 π, while simultaneously and independently controlling β, one can produce also a 

desired fundamental grid power factor (leading, unity or lagging). The active harmonic 

compensator shown in Figure 1 is introduced to further improve the harmonic content and - 

because of the control of the reactive power achieved through β - it does not need to provide any 

reactive power at fundamental frequency. 

Table 1. Values of the Phase-Control Angle φ Reducing the Harmonic Content of the Grid Line 

Current 

φ = ( ) /109 1 π (162°) Nullifies the 5th harmonic 

φ = ( ) 14 / 13 2 π (167.1°) Nullifies the 7th harmonic 

φ = 0.914 3 π (164.52°) Minimizes the total average power in the 5th and 7th harmonics 
6 

Figure 4. Amplitudes of the 5th and 7th harmonics versus the phase-control angle

The capability of controlling the generated reactive power is very advantageous for distributed 

power generation systems such as wind turbines. Neglecting the CSI losses, the average power at 

the grid is equal to the average power on the DC side, <Vdc>·Idc , and Equation (3) shows how the 

average value of the sum of the DC-link bridge voltages, <Vdc>, depends on α and β, thereby 

providing the capability of controlling Idc . Figures 5a and 5b show two simulated waveforms for 

ia with α = 167° and with β = 40° leading and β = 0°, respectively. 

0 90 180 270 360

α = 167°, β = 40°

φ1 φ2

Vab (secondary side)

<Vdc>

Van (mains)

I

a

 (mains)

0 90 180 270 360

α = 167°, β = 0°

φ1 φ2

Vab (secondary side)

<Vdc>

Van (mains)

I

a

 (mains)

 (a) α = 167°, β = 40° leading (b) α = 167°, β = 0°

Figure 5. Simulated converter waveforms (bottom) for α = 167°. The waveforms in the upper plots 

are bridge 1 DC-link voltage (blue), bridge 2 DC-link voltage (red), Vdc (magenta), and average 

total DC-link voltage <Vdc> (green) 

In both cases, the difference between φ1 and φ2 (2·α) does not change; thus, the shape of the grid 

current waveform remains unaltered, while it is shifted with respect to the grid phase voltage by 

the control angle β. 

The DC-link current can remain bounded in time if, and only if, the rectified generator voltage 

and the sum of the two DC-side voltages of the current-source bridges possess identical average 

values. Because of the nature of the system, no direct or fast-acting control of the generator 
7 

voltage is possible, so that modifying the phase-control angle constitutes the only method by 

which such equality can be achieved. The value of the phase-control angle able to equate the 

aforementioned average voltages is obtained via a proportional integral (PI) regulator whose 

input is the error between the desired and actual DC-link current. It can be observed that the DClink current contains all the information necessary about the equality between those average 

voltage values. When they tend to differ, a diverging trend appears in which the current and the 

PI controller act to eliminate this error by modifying the phase-control angle properly. It is then 

sufficient, in principle, to measure only the DC-link current at the base of the tower, thereby 

avoiding other sensors in the nacelle and the consequent complexity and cost to transmit their 

measurement to the controller located at the ground level. With this control strategy, the rectified 

generator voltage acts like a disturbance as depicted in Figure 6. Nonetheless, it must be pointed 

out that for some combinations of DC-link inductance, PI parameters, and control performance 

requirements, it might be worthwhile to also measure the rectified generator voltage to provide a 

feed-forward action able to improve the control loop dynamic. This measurement is present in 

the prototype, although it is not presently used. 

Additionally, it must be noted that the blade-pitch mechanical control of a wind turbine acts 

inside a good range of wind speeds to stabilize the permanent magnet generator voltage at its 

nominal value characterized by the desired rated power. It is then advisable to design the 

generator nominal voltage so that its average value is as close as possible to the total average 

value of the two current-source bridges in series when they operate with the optimum phasecontrol angle 0.914 π. This design choice greatly contributes to the reduction of the 5th and 7th 

harmonics of the grid line current at rated power, already without the correction performed by 

the active harmonic compensator. The phase-control angle is clearly upper value bounded, 

mainly by the necessary recovery time in case of use of a thyristor bridge. Should large sudden 

grid voltage drops and/or voltage escalations in the generator occur, it might become impossible 

to limit the DC-link current without exceeding the maximum allowed value for the control angle. 

In this case, the controller provides a proper overcurrent emergency protection to keep the DClink current at safe values. It is also desirable to detect the onset of discontinuous current 

operation in the DC link, which could occur at very light load conditions. In such cases, it would 

be advantageous to simply disconnect the entire system from the grid since the power generated 

under this condition is minimal. The operational principle of the control loop implementing this 

control task is shown in Figure 6 for the case of β = 0; i.e., unity fundamental power factor at the 

grid under all operating conditions. 
8 

Figure 6. Operational principle characterizing the controller of the main conversion chain for β = 0 

Control of the Active Harmonic Compensator

The second control task consists of synthesis of the proper corrective currents that the active 

harmonic compensator (a classical voltage source inverter [VSI] three-phase inverter) must inject 

in the main conversion chain to achieve a further reduction of the harmonics in the grid line 

currents. Such a task can be conceptually subdivided in three sequential steps listed below and 

illustrated in Figure 7. For the sake of cost reduction, it is preferable to connect the VSI at one of 

the transformer secondaries, although in principle it could operate also at the primary. The 

control task includes: 

1. Estimating the amplitude that the purely sinusoidal reference grid line currents must possess 

to deliver the same average power at the grid 

2. Using such grid reference currents to compute indirectly the specific reference currents for 

the active harmonic compensator 

3. Tracking and synthesizing these last currents through a control technique based on the realtime minimization of a proper time-varying functional algorithm. 

The required amplitude of the desired purely sinusoidal grid reference current is obtainable 

through the simultaneous measurement of the average power delivered to the grid and the 

amplitude of the grid phase voltages. Because the fundamental power factor is already 

guaranteed to be unity (or any desired value lagging or leading) by the previously illustrated 

control technique used for the main conversion chain, a simple algebraic division suffices in 

obtaining the required current amplitude. It must be observed that the measurement of the 

average power is insensible to the harmonic contents in the transformer primary line currents. By 

exploiting this property, the aforementioned amplitude estimation is decoupled from the action 

of the active harmonic compensator. The amplitude estimated at this point must still be corrected 

by the contribution of the PI regulator devoted to maintain the VSI DC-link voltage close to its 

desired value. Indeed, if the VSI DC-link voltage decreases or increases, the PI regulator 

increments or depletes the estimated amplitude respectively. 
9 

Figure 7. Operational principle characterizing the controller for the active harmonic compensator 

In this manner, a quota of energy is delivered or withdrawn from the VSI, thereby stabilizing its 

DC-link voltage. Once the correct amplitude of the reference grid currents becomes available, 

the corresponding sinusoidal waveforms for the line currents are straightforwardly obtained 

through normalization of the measured phase voltages and consequent multiplication by the 

estimated amplitude. At this point, where the reference waveforms for the grid line currents are 

known, it is possible to compute indirectly the specific corrective reference line currents to be 

synthesized by the VSI when it is connected to the same secondary of the fully controllable 

current-source bridge. This step is quite straightforward and involves only algebraic operations, 

provided the structure and transformation ratios of the transformer are known, together with the 

measured line currents at the outputs of the two current-source bridges. The last step is the 

effective tracking of these corrective reference currents performed by the VSI via proper 

switching commands issued to its fully controllable (usually insulated gate bipolar transistor 

[IGBT]) switches. The details of the VSI current-tracking controller have been published by the 

investigators in [1]. 

It is nevertheless important to note the basic principle on which such controller is based. The 

control exploits the real-time minimization of a proper time-varying functional, which, 

converging to its minimum, gives the optimal switching states for current-tracking. Such 

functional represents the “distance” among the actual and reference waveforms, as defined in the 

branch of mathematics usually named “functional analysis.” The minimization of the functional 

is performed on a discrete-time basis defined by a clock signal (whose frequency is not greater 

than 10 kHz to limit the switching losses in the semiconductors), and the overall resulting 

controller action could be classified as a type of carrierless PWM. Because the clock frequency 

will always be limited in practice, a reduced-size L-C filter is advisable at the grid side to filter 

the remaining high-frequency harmonics. 

The following simulation results obtained from the joint operation of Simplorer and Simulink are 

provided to illustrate the operation of the two control tasks previously explained. Figure 8 shows 

the grid line currents (blue) before 0.25 s, when only the two current-source bridges are active, 

and after that instant when the harmonic compensator is enabled. Figure 9 shows the harmonic 

spectrum (computed after 0.25 s.) up to 2 kHz and shows that no harmonic exceeds 5.2% of the 

fundamental. Although before 0.25 s. the quality of such waveforms appears poorer, they possess 

the minimum 5th and 7th harmonics thanks to the overall choices explained above and detailed in 
10 

the following. Figure 10 shows the voltages of the two bridges (blue and red) and their sum 

(magenta), which, together with the rectified generator voltage (green), influences the DC-link 

current. Figure 11 shows the regulation performed by the DC-link controller, which stabilizes the 

phase-control angle close to the optimum value of 165° also after the intervention of the 

harmonic compensator. This highlights the fault-tolerant decoupling between the two controllers. 

Figure 12 shows the instantaneous power (negative) withdrawn from the grid, while Figure 13 

depicts the shape of the generator phase electromagnetic frequency (EMF) originally assumed 

for the permanent magnet (PM) generator. Figure 14 compares the estimated (magenta) and 

actual (blue) Grid line currents, while Figure 15 illustrates how the “minimum distance” 

controller (operating at 10 kHz clock frequency) tracks the corrective line currents (magenta) 

demanded by the controller to improve the spectrum of the grid line currents. 

Figure 8. Grid line currents (blue) corrected by the VSI 
11 

Figure 9. Spectrum of the corrected grid line currents 

Figure 10. Voltages influencing the DC-link current 
12 

Figure 11. Regulation performed by the DC-link controller 

Figure 12. Instantaneous power delivered to the grid 
13 

Figure 13. Simulated generator phase voltages 

Figure 14. Reference and actual grid line currents 
14 

Figure 15. Reference and actual VSI line currents 

Structure and Primary Features of the PM Axial Flux Generator

The structure and related drawings of the final PM generator constructed by McCleer Power and 

Light Engineering are reported in Figure 16, and the waveforms of two line-to-line back-EMFs 

provided by the manufacturer are shown in Figure 17. The primary features of the machine are as 

follows: 

• 48 poles 

• 19-inch external diameter 

• 539 V line-to-line nominal root mean squared (RMS) voltage at 700 rpm nominal speed 

• Two-sided stator, with a single rotor disk in the middle 

• Uses amorphous steel laminations with open slots so that terminal inductance is very low 

• Phase synchronous inductance of about 2 mH and phase leakage inductance of about 1.2 

mH 

• Phase winding resistance of about 0.58 Ω. 

The short-circuit current at 700 rpm is approximately 90 A RMS, which is actually slightly larger 

than the latest 6-pole design by Wisconsin Electric Machines and Power Electronics Consortium 

(WEMPEC). The major external difference from such a design is the operating frequency, 280 

Hz instead of 35 Hz, which leads to a rectified fundamental frequency of 1440 Hz instead of 210 

Hz. This much higher frequency is still acceptable to avoid DC components in the 

interconnection transformer. Because of the much smaller fundamental period, the main 

drawback lies in the higher influence of the commutations in the rectifier, which occurs despite 

the low leakage inductance. To minimize this effect, fast diodes have been used in the prototype 

in which the diode rectifier bridge is obtained by using the antiparallel diodes of three 1200 V 

Semikron IGBT modules, one for each phase leg. 
15 

Figure 16. Mechanical drawings of the PM generator frame 

Figure 17. Waveforms of two line-to-line back-EMFs
16 

Overview and Features of the Chosen DSP-Based Control System

This section provides a short description of the final digital signal processing (DSP) system 

chosen for this research. The controller is a XC 2000 DSP manufactured by AixControl GmbH 

in Aachen, Germany. Figure 18 shows the portable frame (19-inch rack form factor) that 

contains the entire DSP system including the controller board XCS 2000, the interface boards, 

the communication board, and the power supply for all of these components. 

The double-processor controller board XCS2000 is shown in Figure 19. The manufacturer has 

customized the board for this project by providing 24 analog input channels, 8 analog output 

channels, and 32 digital input/output channels. The board is based on two Analog Devices, 

SHARC 21161N DSPs programmable through the Analog Devices Visual DSP ++ crosscompiler. The manufacturer provides libraries in C and C++ specifically oriented for power 

electronics control tasks. The board also features an optoisolated RS-232 interface. 

Figure 18. Self-contained frame hosting the XC 

2000 system

Figure 19. XCS 2000 controller board

Four signal conditioning and interface boards complement the XCS 2000. Three of these are 

identical XCI 2000 boards shown in Figure 20 and will be used to control the three bridges 

present in the topology: two boards for the two CSI bridges and one for the VSI harmonic 

compensator. The XCI 2000 allows the measurement of eight analog channels with a dynamic 

range of ±10 V. protected up to ±40 V., while it simultaneously provides the eight independent 

control signals for four independent standard phase legs. A particularly remarkable feature of this 

board is its ability to interface itself and directly supply the Semikron Skiip modules, including 

also the analog measurements from their embedded current and voltage sensors. 

The two CSI bridges of the prototype are based on salvaged Skiip modules, which has saved a 

considerable amount of time and cost. However, the project budget could not allow for the 

purchase of a third Skiip module (in excess of $3000) for the VSI harmonic compensator 

identified in Figure 1, which has been manually built. The fourth signal conditioning board, the 

XCI 2020 shown in Figure 21, is used mainly to monitor up to eight analog internal variables of 

the DSP board and provides eight digital outputs. The dynamic range of the XCI 2020 analog 

outputs is ±10 V, and all logic outputs are at transistor-transistor logic (TTL) level, buffered and 

protected against electrostatic discharges. The system also includes a 100 Mbit Ethernet 

communication board (XCC 2000) that provides the ability to transfer acquired data and control 

signals to and from the personal computer (PC) and LabView. 
17 

Figure 20. XCI 2000 four-phase converter 

control card

Figure 21. XCI 2020 universal input/output 

card

Prototype and Experimental Results

Figures 22 through 33 show views of the 10-kW experimental setup built in the WEMPEC 

laboratory. 
18 

Figure 22. Diagonal view of the experimental setup with the Variac and drive frame in the 

foreground
19 

Figure 23. Rear view showing the two CSI bridges, DSP controller, and some of the voltage and 

current probes connected to the instrumentation
20 

Figure 24. Diagonal view of the setup showing the dynamometer and the PM axial flux generator
21 

Figure 25. Detail of the control panel for the entire prototype. The PC is used to program and set 

the references to the DSP controller, as well as to collect the measurement from the oscilloscopes

Each bidirectional voltage blocking switch in the prototype has been realized by the series 

combination of one IGBT (with integrated anti-parallel diode) and one external diode of equal 

voltage and current rating (1200 V and at least 100 A). 
22 

Figure 26. The interconnection transformer, axial flux PM generator, and dynamometer (prime 

mover emulating the propeller of the wind turbine)
23 

Figure 27. Drive for the dynamometer
24 

Figure 28. PM axial flux generator (left) coupled with the dynamometer (right)
25 

Figure 29. The PM axial flux generator built by Light Engineering

. 
26 

Figure 30. Diode bridge (bottom center) together with the contactors and fuses for DC-link and 

grid
27 

Figure 31. Protection circuit for the DC-link inductor described in [3] and reported in the Appendix

Figure 32. P. Tenca machining parts on the center lathe
28 

Figure 33. P. Tenca machining parts on the milling machine

A selection of experimental results summarizing the control action on the main conversion chain 

is shown in Figure 34 (a-d) where the operating conditions are specified beneath each plot. These 

experimental results show the grid phase voltage Va (magenta, 50 V/div.) and grid line current ia

(yellow, 20 A/div. for Figure 34a-c), 10 A/div. for Figure 34d) for various values of β, with α

close to the analytically derived value which minimizes the 5th and 7th harmonics (164.52°). The 

Fast Fourier Transform (FFT) of the line current ia is shown in orange in the lower and dedicated 

systems of axes (5·105

 samples, rectangular window, 60 Hz/div. horizontal, vertical scale = 1⁄2 

time domain scale per div.). 
29 

Figure 34a. α = 167°, β = 0°, Vll = 122VRMS, <Vdc> = 325 V, Idc = 15 A

 

Figure 34b. α = 167°, β = 40° leading, Vll = 120VRMS, <Vdc> = 248 V, Idc = 15 A
30 

Figure 34c. α = 167°, β = 40° lagging, Vll = 120VRMS, <Vdc> = 248V, Idc = 15 A
31 

Figure 34d. α = 169°, β = 89.9° leading, Vll = 80.8VRMS, <Vdc> = 10.4 V, Idc = 10 A (10 A/div.)

 

In Figure 34 (a-d) one can see that the 5th and 7th harmonics maintain a very reduced amplitude 

while the current waveform is shifted by the imposed angle β that can reach values very close to 

π/2, also leading, without significant distortion in the grid current waveform. 

Further results, reported in Figures 35 through 42, are aimed at confirming the nullification of 

the 5th or 7th harmonics for different values of the angle α (with β = 0) as predicted by Equation 

(1) and Figures 3 and 4. Suitable comments are inserted directly on the captions. The average 

value of the DC-link voltage for these traces was about 160 V and Idc = 10 A (10 A /div.). 

Figures 43 through 47 show additional results when the angle β is controlled while keeping α at 

167°. 
32 

Figure 35. α = 167.8°. Zero of 7th harmonic
33 

Figure 36. α = 167°. 5th and 7th harmonics minimized

Figure 37. α = 164.4°. Zero of 5th harmonic
34 

Figure 38. α = 143.9°. Zero of 7th harmonic

Figure 39. α = 129.5°. Zero of 5th harmonic
35 

Figure 40. α = 120°. Zero of 7th harmonic

Figure 41. α = 111°. 5th harmonic greater than the 7th
36 

Figure 42. α = 94.7°. The grid phase current is negligible

Figure 43. β = 30°. PF leading
37 

Figure 44. β = -30°. PF lagging

Figure 45. β = 60°. PF leading
38 

Figure 46. β = -64.6°. PF lagging

Figure 47. β = 89.9°. PF leading
39 

Figure 47 shows a transient response for a step in the DC-link current command from 1 A to 10 

A, while Figure 48 shows the opposite transient from 10 A to 1A. The other operating conditions 

are the same. The yellow waveform is the grid current (20 A/div.) while the magenta waveform 

is the DC-link current (2 A/div.). 

Figure 48. Transient from 1 A to 10 A of DC-link current at <Vdc> = 328 V, 1464 rpm

Figure 49. Transient from 10 A to 1 A of DC-link current at <Vdc> = 328 V, 1464 rpm
40 

Suggestions for Future Work

The investigators suggest that the DSP-implementation and the test of the variational control 

method described in this text be applied to the active harmonic compensator. This method could 

be applied in a more general context for current control of any type of power electronic power 

converter. 

Conclusions

The proposed line-frequency switched CSI topology allows the use of high-power, slower 

devices, making it suitable for high-power applications. This topology is particularly well suited 

for wind-power applications as only the generator and rectifier bridge (consisting of very reliable 

diode semiconductors) are located in the nacelle, and the two DC link cables that run from the 

nacelle to the ground can contribute to a significant portion of the DC link inductance. The axial 

flux PM generator allows the realization of a gearless turbine and greatly improves the reliability 

of the generator because it does not require external or brushless excitation. With the proposed 

control scheme, the topology can simultaneously provide relatively low line current harmonics at 

the grid while producing either leading, unity, or lagging power factor. These features are 

already achieved without using the active harmonic compensator. The possible use of a harmonic 

compensator would decrease the harmonic content further and, because control of the reactive 

power at fundamental frequency is provided by the control angle β, it needs to be designed and 

rated only for the harmonic components above the fundamental component. 

Published Papers Related to the Project Activities

1. Tenca, P.; Lipo, T.A. “Reduced cost current-source topology improving the harmonic 

spectrum through on-line functional minimization.” Power Electronics Specialists Conference, 

2004. Vol. 4, 2004; pp. 2829-2835. 

2. Tenca, P.; Lipo, T.A.“Synthesis of desired AC line currents in current-sourced DC-AC 

converters.” Second International Conference on Power Electronics, Machines and Drives, 

March 31 - April 2, 2004. Conf. Publ. No. 498., Vol. 2; pp. 656-661.

3. Tenca, P.; Lipo, T.A. “A decentralized protection scheme for converters utilizing a DC-link 

inductor.” Industrial Electronics Society, 2005. IECON 2005. 32nd Annual Conference of IEEE 

November 6-10, 2005; pp. 854 – 859 (to appear in IEEE Transactions on Industrial Electronics). 

4. Tenca, P.; Lipo, T.A. “Conversion Topology for Reducing Failure Rate and Life-cycle Costs 

of High-Power Wind Turbines.” ASME-AIAA Wind Energy Symposium, pp. 111-122, 2005. 
41 

Appendix

Publication [3] in the above list of references refers to a passive circuit developed to protect the 

DC-link inductor against dangerous overvoltages and/or overcurrents potentially occurring after 

a fault. This circuit was not specified in the original proposal but was developed as a portion of 

this project as an additional effort. Because the description of such a circuit would have diverged 

quite considerably from the main topic of this report, it is included here as an appendix. 
42 

A Decentralized Protection Scheme for Converters Utilizing a DC-link 

Inductor 

Pierluigi Tenca Thomas A. Lipo

University of Wisconsin-Madison University of Wisconsin-Madison

Engineering Hall, 1415 Engineering Drive Engineering Hall, 1415 Engineering Drive

Madison, WI, 53706 U.S.A. Madison, WI, 53706 U.S.A.

tenca@cae.wisc.edu lipo@engr.wisc.edu

Abstract – This paper proposes a fully passive protection scheme for DC link inductors whose topology is characterized by 

the circuit having a substantial distance between the rectifier and inverter stations. Inductors of this type can be found 

intrinsically in very long high-power cables - as in HVDC transmission for example - as well as in DC-links for highpower current source inverters. In such applications the inductor must be duly protected against overvoltages as well as 

against overcurrents potentially occurring as consequence of an open-circuit or a short-circuit fault respectively. The 

distance between the inductor terminals prevents the use of well-known localized protections like, for example, simple 

freewheeling diodes. A key role in the protection scheme proposed by the authors is played by the largely available 

transient suppressor diodes (often commercially identified as “Tranzorbs”) and the solution is kept very simple and 

reliable by employing only rugged diodes and thyristors. The protection relies only on the energy stored in the inductor 

without the need of any external supervision circuit or any information exchange between the distant inductor terminals. 

Experimental results are presented and a generalization of the scheme to a broader application is also introduced. 

I. INTRODUCTION 

Often it is necessary to protect the insulation and mechanical integrity of large DC link inductors from damage 

caused by excessive overvoltages and/or overcurrents usually arising from open-circuit or short-circuit faults 

respectively. Such large inductors are a very expensive and important parts of the systems in which they operate. For 

this reason the task of designing a proper protection scheme for them must not be underestimated. 

Often large DC current link applications are characterised by the fact that their terminals are located at very 

considerable distance apart (even several kilometers) as Figure 1 shows. The most common practical case occurs 

when the effective distribution of the inductance is over long distance, as may occur for high-power transmission 

cables or for inductors used as DC-link in high power current source inverters / rectifiers [1]. 

The inductors in Figure 1 have been drawn to call attention to the distributed nature of their inductance which is 

always a property of the overall loop in which the inductor current flows. Often the inductor constitutes the DC-link 

of current source converters in which the current iL as well as the voltages vA and vB do not change sign. 

Fig. 1. Situation in which the effective terminals of a DC link inductor are distant from each other. 

The distributed nature of the inductance leads one to approach the problem of protection of such inductors in a 

decentralized manner. Firstly, it is useful to observe that ultimately the portion of electrical network needing 

protection is the quadrupole A1-A2-B1-B2 identified in Figure 1 and not only a simple bipole. 

As consequence the common protection against open-circuit faults constituted by a simple freewheeling diode can 

not be applied effectively. Indeed it is sufficient to observe that the effectiveness of two possible freewheeling 
43 

diodes, one connected between the terminals A1 and B1 and the other between A2 and B2, would be jeopardized by 

the unavoidable large inductances of the diodes connections to A1 (A2) and B1 (B2). The magnitude of such 

undesired inductances would arise naturally as consequence of the large distance between the converter terminals. 

Furthermore a freewheeling diode would not provide any protection against overcurrents or overvoltages possibly 

arising at the ports A1-A2 and B1-B2 of the quadrupole. 

In order to be as more reliable as possible, the protection scheme should be kept as simple as possible, since every 

added complexity introduces an additional risk of malfunction which is less and less tolerable as the cost of the 

inductor increases. In particular the protection scheme should possess the following features: 

- Absence of any external supervision circuit. 

- Independence from any external energy source. 

- Capability to operate using only the energy stored in the inductor. 

- Absence of information exchange between the two distant ports A1-A2 and B1-B2. 

II. PROPOSED PROTECTION SCHEME 

As mentioned, it is common that the inductor represents the DC-link of high-power current source converters with 

the quantities iL , vA and vB having constant sign that can be assumed positive under normal operating conditions. 

Unless stated otherwise these assumptions will be at the basis of the subsequent discussion. 

All quantities iL , vA and vB clearly have design limits and considering the requirements listed in the previous 

paragraph the authors propose the circuit solution shown in Figure 2 as a decentralized protection scheme. The 

symbols FA1, FA2, FB1, FB2 represent fuses that might be fast-acting or slow-acting depending on the application. The 

tripole identified as “Thyristor trigger circuit” senses the voltage vB and triggers the thyristor TB whenever vB 

exceeds the design value considered as the symptom of a fault. 

Observing Figure 2 it is possible to analyze how the protection scheme operates in the different possible fault 

conditions. 

I) An open-circuit fault either occurring in network A or consequence of the intervention of fuses FA1 and/or FA2, 

causes the current iL to flow into the diode DA and the voltage vA to equal the negative forward voltage drop of DA. 

The current iL decreases after the fault because of the action of the voltage vB. 

II) An open-circuit fault either occurring in network B or consequence of the intervention of fuses FB1 and/or FB2 , 

causes the voltage vB to rise until its fault value is reached. At this instant the thyristor TB is ignited through the 

trigger circuit and the terminals B1 and B2 are almost short circuited. 

If the positive voltage vA is not reduced considerably after this event, the current iL raises under its presence until 

at least one of the fuses FA1 and FA2 intervenes, because its time-integral of the current squared has been exceeded. 

In this case the current iL is diverted to flow in DA. 

Fig. 2. Basic principle of decentralized protection scheme. 
44 

The short circuit produced by the thyristor TB at the port B1-B2 ultimately leads to the intervention of at least one 

of the fuses FB1 and FB2 unless additional possible – but not required - protection policies operating on the network B 

prevent this from happening. If this is not the case the ultimate operating mode of the inductor is the one in which 

the current iL may flow in DA and TB only while its value decreases towards zero under the action of the forward 

voltage drops of these devices. 

III) Any faulty operating condition creating an overcurrent iL which exceeds the I2

t rating of the fuses will 

ultimately melt at least one of them. As consequence, from that instant onwards the protection scheme behaves like 

one of the previous cases I or II. 

The scheme provides also fast protection for the inductor against overvoltages at port B1-B2 because such an 

event invokes the intervention of the thyristor trigger circuit and the consequent behaviour as described by the 

previous case II. On the other hand an overvoltage condition at the port A1-A2 is detected indirectly and with slower 

dynamics because it must create a rise of the current iL or - if a controls system for iL is present in the network B - an 

elevation of the voltage vB. 

This differentiation between the mode of intervention for overvoltages in vA and vB is justified by the hypothesis 

under which this principle scheme has been developed. Indeed when the inductor constitutes the DC-link of a 

current source converter operating with positive iL, vA and vB, as assumed, the network A, which is the generating 

side, may be more affected by short spike voltages that should be ignored. 

It should be observed that the proposed scheme strongly favors the protection of the inductor over the networks A 

and B. This choice arises from the original assumption that the inductor is a highly valuable component whose 

integrity and survivability deserve the highest priority. 

III. STRUCTURE OF THE THYRISTOR 

TRIGGER CIRCUIT 

Thus far only the function of the thyristor trigger circuit has been addressed but not its detailed structure and 

accompanying design choices. First of all it should be observed that when the occurring faulty condition ultimately 

leads to the operating mode II described in the previous section, the inductor current iL cannot be diverted 

instantaneously into the thyristor TB because of its finite commutation time. Consequently the thyristor trigger 

circuit must be able to withstand the entire fault value of iL until TB goes in full conduction. 

Additionally, in general, it is not possible to identify a single value of iL that could be used as decisive threshold 

for triggering the thyristor. This is because an open circuit fault in network B could occur for reasons completely 

independent from iL and its value. These observations together with the requirements listed at the end of section I 

restrict the set of possible components that can be used in the trigger circuit. 

The authors have adopted the structure and components shown in Figure 3 to realize the trigger circuit enclosed in 

the red rectangle for sake of clarity. In order to satisfy the requirement of operating only with the energy stored in 

the protected inductor, the circuit employs only passive components and does not require any external energy 

source. 

It should be further observed that such a feature is very important because it assures protection even in case of 

catastrophic fault of the entire system, with consequent loss of all energy sources. In a scenario like this, one can 

rely only on the energy stored in the inductor. The trigger circuit uses only the local information represented by the 

voltage vB and does not require other additional external supervision circuits. 

The key detail for the correct circuit operation is the utilization of the nowadays widely available unidirectional 

Transient Voltage Suppressors (TVS) diodes which are often denominated with the conventional name of 

“Tranzorb”. These components clamp the voltage to a maximum value close to their nominal one with effects 

similar to a zener diode. Nonetheless, very differently from the zener diodes, they can withstand and dissipate very 

high instantaneous powers for few microseconds and, furthermore, their first failure mode is guaranteed to be a short 

circuit [2]. 

Tranzorbs rated for 1.5 kW – 3kW of dissipated power are commonly produced by many manufacturers and 

nowadays devices rated for 30 kW and 6 kA peak current are emerging produced, for example, by Sussex 

Semiconductors [3]. The manufacturer usually specifies the waveform of the surge current that a Tranzorb can 

withstand safely as an asymmetrical bell-shaped curve having a rise time of 10 μs and a fall time, to 1⁄2 of the peak 

current, of 1000 μs. 
45 

Fig. 3. Structure of the fully passive thyristor trigger circuit. 

The nominal peak surge current is roughly equal to the rated power of the device divided by its nominal clamping 

voltage. The Tranzorbs are widely used as protection devices against overvoltages (especially induced by lightning) 

since they can clamp the voltage much more effectively than Varistors [4]-[5] and they are guaranteed to fail as a 

short circuit providing an ultimate protection that is generally used to blow a fast-acting fuse. 

The features of the Tranzorbs are particularly suitable for the thyristor trigger circuit because they are components 

of reduced size which are nevertheless able to withstand a large impulse current. This is exactly the situation 

occurring in the trigger circuit inside the time interval between the inception of the fault and the instant when TB 

goes in full conduction. In this interval the surge current iTZ in the Tranzorbs becomes higher than iL and the devices 

must be chosen so that their peak current is not lower than the maximum iL that can occur in the worst case fault 

condition. Zener diodes could not be used as substitute of Tranzorbs because there is no guarantee on their clamp 

voltage and failure mode for peak currents well above their absolute maximum rating. 

Following these considerations, it can be observed in Figure 3 that the branch composed by the series of lowvalue resistor Rz, Tranzorb Tzf and n Tranzorbs Tz_1 – Tz_n, constitutes a path in which the inductor current iL can 

flow - until TB reaches the full conduction - without risk of being abruptly interrupted with consequent onset of 

potentially destructive values of vB. The number of Tranzorb diodes in series implies the maximum overvoltage for 

vB after which the circuit ignites TB. As soon as a specific Tranzorb device has been selected according to additional 

criterion, such as availability for example, the desired maximum vB determines the number n+1 of devices needed in 

series. 

The layout of the trigger circuit must be as compact as possible and lie as close as possible to the port B1-B2 in 

order to minimize the parasitic series inductance on the path of iL. To relax in some extent such a requirement and to 

protect TB against excessive dvB / dt the very low series impedance capacitor CB is introduced and it is mounted as 

close as possible to the TB terminals. 

From Figure 3 one can infer the succession of events which lead TB into conduction after an open circuit fault of 

type II. Before the fault inception the voltage vB has a positive value lower than the threshold defined by the number 

of Tranzorbs in series and their avalanche voltage. As soon as the open circuit fault occurs the current iL flows into 

the capacitor CB and the voltage vB raises linearly up to the maximum clamping voltage defined by the series of n 

Tranzorbs Tz_1 – Tz_n. At such instant these Tranzorbs conduct and the voltage vB remains almost clamped. As 

consequence, the current iTZ becomes equal to iL because the capacitor voltage is practically constant. Not later than 

when the voltage of the small filtering capacitor Cf reaches the clamping voltage of the Tranzorb Tf, a gate current 

ig appears and its peak is set by the difference of the clamping voltages of Tzf and Tg as well as the resistor Rg. As 

soon as ig exceeds the thyristor gate trigger current, TB reaches the full conduction after the delay time specified by 

the manufacturer which is usually below 5 μs. At this point the thyristor current iS starts to rise with a slope 
46 

determined by the clamping voltage of the series of Tranzorbs and the turn-on snubber inductance Ls. The inductor 

Ls must always be present to limit the maximum di/dt for which the thyristor is rated. Ultimately iS will become 

equal to iL, iTZ will drop to zero ending so the commutation process and the surge current in the Tranzorbs. It is 

important to choose a thyristor whose I2

t is greater than the one characterizing the fuses chosen. By selecting 

properly the components the whole area of the pulse current in the Tranzorbs remains easily inside the safe limits 

specified by the manufacturer. 

IV. EXPERIMENTAL SETUP AND RESULTS 

An experimental setup has been built to test the effectiveness of the proposed protection circuit. The setup is part 

of a wider one aimed at testing topologies for high power wind turbines. Figures 4 and 5 show the portion of the 

setup regrouping the power semiconductors and thyristor trigger circuit while the 67 mH inductor to be protected is 

not shown because collocated far away from the other components. 

The components shown in Figure 2 and 3 are highlighted in figures 4 and 5 for sake of explanation and clarity. 

The capacitor CB is composed by one extremely low inductance 470 nF snubber capacitor mounted directly on the 

terminals of the thyristor TB together with additional two 1nF and 600 pF capacitors mounted on the trigger circuit 

board. The Tranzorb Tz_1 – Tz_n and Tzf used are devices 1N6284A (1.5KE36A) rated for 1.5kW and 36 V 

breakdown voltage while the Tranzorb Tg is a 1N6272A (1.5KE11A) rated for 11.5 kW and 11 V breakdown 

voltage. 

Fig. 4. Particular of the experimental setup. 
47 

Fig. 5. Detail of the thyristor trigger circuit showing the components. 

 On the trigger board shown in Figure 5 there are two groups of 11 Tranzorbs each, all connected in series, for a 

total possible breakdown voltage of 792V. In the test performed only one group has been used setting so the 

intervention voltage of the trigger circuit was close to 400 V. The resistor Rf and the capacitor Cf have values of 100 

Ω and 0.1 μF respectively, both selected for low inductance. The resistors Rg and Rz are 22 Ω and 1 Ω respectively, 

again both are antiinductive. The thyristor TB is a device TD310N from Eupec and the turn-on snubber inductance 

Ls used has a value of 30 μH to limit the di/dt at 13.4 A/μs (with 400V maximum VB voltage) well below the 120 

A/μs which is the device limit. 

Two LeCroy digital scopes have been used to perform the measurements. The first scope, a model LT354, (four 

channels, 500 MHz) is used to monitor the waveforms of the trigger circuit vTR (CH1,blue), iTZ (CH2,green), Vf 

(CH3, red) and ig (CH4, magenta) and a capture of its screen during a test with iL = 10 A is shown in Figure 6. The 

second scope, a model 9304 (four channels, 175 MHz) monitors the waveforms vA (CH1, magenta), iL (CH2,blue), 

vB (CH3, red) and iS (CH4, green). The two scopes were connected with the 9304 acting as trigger master on the 

voltage vB and the LT354 as trigger slave 

Fig. 6. Display of the LT354 scope with waveforms of the trigger circuit. 

Two tests were performed on the experimental setup and in both the network A was constituted by a six-pulse 

diode rectifier whose input was a 60 Hz three-phase symmetrical system of sinusoidal voltages regulated in 

amplitude by a Variac. The network B was a 16 Ω power resistor. Between the network A and the fuses FA1, FA2, as 

well as between the fuses FB1,FB2 and the network B, solid state switches (high-power IGBTs) were inserted to 

realize very “clean” open-circuit fault conditions. 

The first test was performed by creating an open-circuit condition at the port B1-B2 after that the inductor current 

iL had reached the steady state value of 10 A. The scopes were triggered for a 200V value with positive slope sensed 

on the voltage vB and the waveforms recorded on a time base scale of 5 μs/div are shown in Figure 7. The upper 

graph of Figure 7 presents the waveforms of vA, iL, vB, iS recorded by the scope 9304 while the waveforms of vTR, 

iTZ, Vf, ig recorded by the scope LT354 are presented in the lower graph. 
48 

By analyzing the traces it is possible to observe the operation of the circuit. After the open circuit occurs at the 

port B1-B2 The voltage vB raises linearly starting from about 160 V. When it reaches a value close to 400 V the 

Tranzorbs Tz_1 – Tz_n begin to conduct. Simultaneous and equal peaks occur in the currents iTZ and ig starting at 

+10 μs because the Tranzorb Tg (11 V) allows the flow of the gate current before than Vf reaches the clamping 

value of 36V. 

Fig. 7. Waveforms for the test with 10A of inductor current. 

The thyristor voltage vTR drops abruptly after ig has reached the ignition value. As consequence of the shortcircuit created by TB the capacitor CB begins a discharge thyristor transient current through LS and indeed one can 

observe that the ramping in the current iS due to this event. After LS has been charged, it will perform a partial 

discharge transient also in the series connection of all Tranzorbs until iS assumes the same value of iL. 

During this transient the Tranzorbs are directly polarized acting as normal diodes. The negative peak observable 

in iTZ is due to this process. It is important to observe that the waveform of the current iL does not show any 

appreciable transient, confirming that iL commutates smoothly from the network B to the short circuit created by the 

triggered thyristor TB; i.e. the protection scheme has accomplished its task of preventing abrupt changes in the 

inductor current. 

The second test performed was aimed at testing the effectiveness of the protection in case the inductor current iL

was of limited value, i.e. in the range of tenths of Amperes. Figures 8 and 9 show the waveforms of iL, vB, iS

recorded during this test repeated on two different time base scales. In Figure 8 one can observe that although the 

open circuit fault occurs when iL is small, nevertheless, vB rises and fires the thyristor through the trigger circuit. 
49 

Fig. 8. Waveforms for the test with low inductor current. 

Fig. 9. Test with low inductor current repeated with magnified time base. 

The waveform of iS shows a peak due to the discharge of CB through the thyristor as explained before. 

Subsequently iS decreases by flowing in the directly polarized Tranzorbs until it reaches the same value of iL. The 

peak of iS, its subsequent decrease and the evolution of vB are magnified in Figure 9. The inductor remains always 

connected to the voltage source at the port A1-A2 while the thyristor TB provides the recirculation path for the 

current iL. As consequence the inductor currents iL and iS begin to rise as clearly shown by Figure 8 and ultimately iL

would provoke the intervention of at least one of the fuses FA1, FA2 and the diode DA as explained in section II, 

thereby completing the sequence of events for the protection of the inductor. 

V. GENERALIZED PROTECTION SCHEME 

The protection scheme presented in the previous sections was shown to be valid for unidirectional power flow 

from network A to network B when the current iL as well as the voltages vA and vB assumed only positive values. 

The authors consider it worthwhile to propose a generalization of the scheme able to operate also in the common 

cases where a bidirectional power flow is required by reversing the signs of the voltages vA and vB while iL

continues to remain always positive. 

The generalized protection scheme under these assumptions is shown in Figure 10. The group composed by the 

thyristor TB and its trigger circuit has been replicated at the port A1-A2 but its polarity has been reversed to allow 

the flow of the current iL during a fault condition. The unidirectional voltage blocking capability of the chosen 

Tranzorbs imposes one additional diode (Dta, Dtb) be provided in series to each trigger circuit at both ports. Without 

this diode the voltage at each port could not reverse its polarity because the Tranzorbs would immediately enter the 

forward conduction mode. 

In the worst fault case, during a hypothetical simultaneous open circuit fault at the ports A1-A2 and B1-B2, the 

current iL will flow initially in the capacitors CB of both trigger circuits until sufficient capacitor voltages are reached 
50 

to ignite the thyristors TA and TB. It is worthwhile to note that the capacitor voltages rise independently and 

simultaneously because the path provided by them is the only one in which iL can flow initially. 

As consequence TA and TB are eventually ignited, but always independently, and still without the need of any 

communication between the circuits located at the two ports. As well it is possible to observe how this protection 

scheme provides overvoltage protection of both signs although each specific port is capable of sensing only one 

specific voltage polarity. 

Fig. 10. Decentralized protection scheme for bidirectional power flow. 

VI. CONCLUSIONS 

A decentralized protection scheme designed to protect geometrically large DC-link inductors from overvoltages 

and overcurrents has been presented. The protection of the DC-link inductor is considered as a priority choice in 

case of fault. The proposed protection scheme does not require any supervision circuit or communications between 

the distant terminals of the inductors to be protected and uses passive devices only, without any need of external 

power sources, as it exploits the same energy stored in the inductor before the fault event. An experimental setup has 

been realized to test the effectiveness of the protection scheme and the related results have been presented. 

Additionally, a possible generalization of the scheme capable to operate also in case of bidirectional power flow 

across the DC-link inductor has been proposed. 

VII. REFERENCES 

[1] P. Tenca, T. A. Lipo, "Reduced cost current-source topology improving the harmonic spectrum through on-line 

functional minimization," in Proceedings of the 2004 IEEE 35th Annual Power Electronics Specialists Conference 

(PESC 2004), Volume 4, pp. 2829-2835. 

[2] M.O. Durham, K.D. Durham, R.A. Durham, “TVSS designs,” IEEE Industry Applications Magazine, vol. 8, no. 

5, Sept-Oct. 2002, pp. 31-36. 

[5] Sussex Semiconductor Products Listing, available at www.sussexsemiconductor.com 

[4] J.M. Diamond, “Varistor control of inductive transients," IEEE Trans. on Circuits and Systems I, vol. 39, no. 6, 

June. 1992, pp. 478-480. 

[5] M. Bartkowiak, M.G. Comber, G.D. Mahan, “Failure modes and energy absorption capability of ZnO varistors” 

IEEE Trans. on Power Delivery, vol. 14, no. 1, Jan. 1999, pp. 152-162. 
F1146-E(12/2004) 

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1. REPORT DATE (DD-MM-YYYY)

July 2006 

2. REPORT TYPE

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3. DATES COVERED (From - To)

October 29, 2002-December 31, 2005 

4. TITLE AND SUBTITLE

Design and Test of a Variable Speed Wind Turbine System 

Employing a Direct Drive Axial Flux Synchronous Generator 

5a. CONTRACT NUMBER

DE-AC36-99-GO10337 

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6. AUTHOR(S)

T.A. Lipo and P. Tenca 

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NREL/SR-500-39715 

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WER6 0301 

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14. ABSTRACT (Maximum 200 Words)

The goal of this funded research project is the definition, analytical investigation, modeling, and prototype realization 

of a current-source conversion topology tailored to high-power wind turbines. 

15. SUBJECT TERMS

wind energy; wind turbine; grid; generator; voltage; converter 

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