NTIA Report 01-383

The Temporal and Spectral

Characteristics of Ultrawideband Signals

William A. Kissick, Editor

U.S. DEPARTMENT OF COMMERCE

Norman Y. Mineta, Secretary

Gregory L. Rohde, Assistant Secretary

for Communications and Information

January 2001
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iii

Disclaimer

Certain equipment and software products are identified in this report to ensure completeness and

accuracy in describing the information presented. Such identification, implied or specific, does

not represent a recommendation or endorsement of the companies or the products by the

National Telecommunications and Information Administration or the National Institute of

Standards and Technology.
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v

CONTENTS

Page

EXECUTIVE SUMMARY .................................................... vii

ABSTRACT ................................................................ 1-1

1. THE RADIO SPECTRUM AND ULTRAWIDEBAND ............................ l-1

2. UWB TECHNOLOGY AND REGULATORY ISSUES ........................... 2-1

3. ANALYTICAL DESCRIPTION OF TIME AND SPECTRAL CHARACTERISTICS

OF ULTRAWIDEBAND SIGNALS .......................................... 3-1

4. CHARACTERISTICS OF AN AGGREGATE OF ULTRAWIDEBAND SIGNALS ..... 4-1

5. FULL-BANDWIDTH REFERENCE MEASUREMENTS OF ULTRAWIDEBAND 

EMISSIONS ............................................................. 5-1

6. BANDWIDTH-LIMITED MEASUREMENTS OF ULTRAWIDEBAND DEVICE 

EMISSIONS ............................................................. 6-1

7. EFFECTS OF TWO UWB SIGNALS ON THREE FEDERAL RADARS ............. 7-1

8. MEASUREMENT SUMMARY AND CONCLUSIONS ........................... 8-1

9. COMPARISON OF THEORY AND MEASUREMENTS .......................... 9-1

APPENDIX A. Amplitude Probability Distributions ............................... A-1

APPENDIX B. Simulations of Time and Spectral Characteristics of 

Ultrawideband Signals and Their Effects on Receivers .............................. B-1

APPENDIX C. Conversion of Power Measured in a Circuit to Incident Field Strength 

and Incident Power Density, and Corrections to Measured Emission Spectra 

for Non-Constant Aperture Measurement Antennas ................................. C-1

APPENDIX D.

D.1 Device A Data ..................................................... D-A-1

D.2 Device B Data ...................................................... D-B-1

D.3 Device C Data ...................................................... D-C-1

D.4 Device D Data ..................................................... D-D-1
vi

D.5 Device E Data ....................................................... D-E-1

D.6 Electric Drill ........................................................ D-F-1

APPENDIX E. Summary of Aggregate Measurements ............................... E-1
vii

EXECUTIVE SUMMARY

Objectives of this Work

A preliminary objective was to develop a description and gain an understanding of the

Ultrawideband (UWB) signal structure based on current, and hopefully typical, UWB system

capabilities and applications. This began with a determination, from specifications and/or direct

measurement, of the salient temporal characteristics of UWB signals that included minimal

descriptions of their modulation schemes for data and/or voice and detailed descriptions of their

pulse shape, width, repetition rate, dithering, and gating characteristics. Then, key fundamental

aspects of UWB signal behavior were derived from first principles. This provided a basis for

identifying what to measure and the effects certain temporal characteristics have on the spectral

characteristics.

The primary objective was to observe and record the temporal and spectral characteristics of

various UWB signals using both highly accurate measurement methods and practical approaches

with commercial off-the-shelf (COTS) test equipment. The measurements are supported by the

theoretical work noted above and confirmed through simulation. Meeting the primary objective

has provided the technical information needed by NTIA to develop policies for use of UWB by

the Federal government and to work with the Federal Communications Commission (FCC) to

develop rules and regulations for UWB emissions. Secondary objectives included the

development and description of reliable and repeatable measurement methods using COTS test

equipment and to measure the effects UWB signals have on several, selected Federal Aviation

Administration (FAA) radar systems.

There are unanswered questions and claims regarding UWB. Some say that the 2 GHz bandwidth

(nominal, based on a pulse width of 1 ns) is ideal in many applications because the already low

total power of a UWB signal is spread so “thinly” that the spectral power density in any

conventional (bandwidth limited) channel is inconsequential. The claim goes further to say that

the signal is similar to Gaussian, or white, noise therefore it is like the background noise any

communications or radar receiver experiences.

UWB Technology and the Radio Spectrum

UWB technology may offer very effective solutions for various communications and sensing

applications; but its uncommon approach of using narrow pulses, or impulses, as a basic signal

structure rather than generating and modulating a sinusoidal carrier results in an unusually wide 

emission bandwidth. Since such a wide signal covers many radio bands and services, the

conditions under which it can operate without causing undue interference must be determined

before UWB systems are allowed to proliferate.
viii

The use of a carrier signal by nearly all existing services that share the radio spectrum helps

ensure that the bandwidth of the emissions of those signals can be kept as narrow as possible for

any given application, i.e. the bandwidth required to transmit the information of interest or

perform the necessary sensing functions. This approach allows for effective and efficient

spectrum management and frequency assignment procedures for sharing of the radio spectrum

among diverse applications and users. Can UWB share the radio spectrum with existing users?

What frequency-related limits such as emission bandwidth and lower frequency limit should be

imposed on UWB signals? Should limits be established for time-related characteristics such as

pulse width and pulse repetition rate (PRR)? If UWB systems proliferate, what are the effects of

the aggregate of independent UWB signals?

Measurements of Ultrawideband Signals

From over twenty UWB devices available to ITS, five were chosen to be fully measured. This

selection represents a sampling of the various UWB signal waveforms in use. This group

included communications and sensing devices that used pulse-position and on/off keying

modulation methods, some did not incorporate pulse dithering, another used relative dither and

yet another used absolute-time-base pulse dithering, one had gated pulse groups.

A very fast transient digitizer was used to capture the individual pulses directly in the time

domain (in some cases a sampling oscilloscope was used) at the output of each device (a

“conducted” measurement) and “in space” as measured by a known antenna (a “radiated”

measurement). Figure ES.1 shows examples; (a) is a narrow impulse about 1.5 ns in length from

one of the UWB devices, and (b) is a longer, very complex pulse shape about 15 ns in length

from a different device. The former occupies about 3.5 GHz of spectrum and the latter about a

fifth or sixth of that. Although some devices generate an impulse like that shown above in

ES.1(a), when radiated by an antenna, the impulse may be changed quite dramatically. Figure

ES.1(c) shows what the pulse shown in (a) is like after being radiated by an antenna designed to

radiate UWB signals.

Measurements of the UWB signal power in various bandwidths were made using spectrum

analyzers and it was determined that the measurement of the signal amplitude probability

distribution (APD) is a very informative measurand. It shows, sometimes in a dramatic way, the

general nature of a UWB source, whether it resembles Gaussian noise or very impulsive noise.

Figure ES.2 shows two APD curves on a Rayleigh probability scale. Curve A, actually a straight

line, represents a signal that is Gaussian distributed; while curve B is the APD for one of the

UWB devices measured. Notice that the signal exceeds about -55 dBm for 1% of the time and 

exceeds -80 dBm for about 12% of the time that it is on. Both signals here are noise-like; the

former, a Gaussian distribution, is a truly random signal and the latter is a highly impulsive

signal.
ix

-10

-5

0

5

10

5 10 15 20

Time (nanoseconds)

Volts

Figure ES.1(b). An example of a long UWB pulse.

-2

-1

0

1

2

5 10

Time (nanoseconds)

Volts

Figure ES.1(a). An example of a short UWB pulse.
x

-0.030

-0.015

0

0.015

0.030

10 20 30

Time (nanoseconds)

Volts

Figure ES.1(c). The shape of the pulse shown in (a) radiated, i.e.

“in space.”

-120

-100

-80

-60

-40

-20

.001 .1 1 5 10 20 35 50 70 80 90 95 99

Single source

Percent Exceeding Ordinate

Power (dBm)

Figure ES.2. APDs for Gaussian noise and a

UWB signal.
xi

Companion Report and Other Investigations

The research, observations, measurements, and analyses presented in this report were performed

at the NTIA Institute for Telecommunication Sciences (ITS), an independent laboratory located

in Boulder, Colorado. The National Institute of Standards and Technology (NIST) Radio

Frequency Technology Division, also located in the same building as ITS in Boulder, performed

some of the measurements reported herein. The NTIA Office of Spectrum Management (OSM)

has used the results of this investigation to examine the options and constraints appropriate for

allowing UWB to share the use the radio spectrum. In their report, a companion to this one,

separation distances are developed for widely accepted receiver protection and interference

criteria. It discusses the operation of UWB under unlicensed and licensed conditions.

A follow-on research effort at ITS has made use of the knowledge of UWB signal characteristics

from this work to develop a test facility to measure the effects of UWB signals on Global

Positioning System (GPS) receivers. The results of this GPS interference investigation will be

published by ITS in a report similar to this one and OSM will use the GPS receiver performance

data to determine the federal government’s position regarding the potential for UWB to share the

spectrum, in particular, the GPS band at 1.5 GHz.
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1The editor, and author of Section 1, is with the Institute for Telecommunication

Sciences, National Telecommunications and Information Administration, U.S. Department of

Commerce, Boulder, CO 80305. 

THE TEMPORAL AND SPECTRAL CHARACTERISTICS OF ULTRAWIDEBAND 

SIGNALS

William A. Kissick, Editor1

Ultrawideband (UWB) technology, useful for both communication and sensing

applications, uses the radio spectrum differently than the vast majority of

radiocommunication technologies. UWB systems make use of narrow pulses and

time-domain signal processing. Questions regarding how these systems, with their

potentially very wide emission bandwidths, might affect the efficient use of the

radio spectrum or cause interference to conventional radio and wireless systems

must be answered before there is any large-scale deployment of UWB systems.

The investigation reported here examined both the temporal and spectral

characteristics of UWB signals, since all radio signals exist in both the time and

frequency domains. The investigation was approached with theoretical analyses,

measurement of actual UWB devices, and computer simulations. The emissions

of several UWB transmitters were measured under controlled, and repeatable,

laboratory conditions. Those measurement methods useful for routine

measurements using commercially-available test equipment were identified. The

characteristics of an aggregate of several UWB signals were examined. An initial

assessment of the effects of UWB signals on several Federal Government systems

was accomplished through field measurements. This report provides a basis for an

assessment of the effects of UWB signals on other communication and radar

systems, the study of the spectrum efficiency of UWB technologies, and the

development of spectrum sharing policies and regulations.

Key words: emissions, aggregate emissions, ultrawideband, UWB, time domain, frequency

domain, radio spectrum, average power, RMS power, peak power, signal strength,

pulse measurements, spectrum measurements.

1. THE RADIO SPECTRUM AND ULTRAWIDEBAND

William A. Kissick1

As the radio spectrum becomes more crowded due to the ever-increasing demand for radio and

wireless communications and for sensing, a wide variety of creative approaches have been 

proposed for allowing more users to share this limited resource. These innovations include new,

digital technologies that permit the same amount of information (e.g., an audio signal) to fit into

increasingly narrower channels as is occurring in the land mobile radio service; or allow much

more information to be transmitted in existing channels as is occurring with high-definition
1-2

television. This investigation is primarily concerned with one such approach called

ultrawideband (UWB) technology and its ability to share the spectrum with existing users. There

are claims that UWB technology, which uses novel signal generating and processing methods,

can use large portions of the already allocated spectrum with minimal or no interference to

existing users due to the very low spectral power density of UWB signals. The assessment of that

claim is critical to decisions regarding the deployment, and potential ubiquitous use, of UWB

devices for both communications and sensing. The radio spectrum, a nondepleting but limited

natural resource, is used to support all radio and wireless services for both public and private

purposes. Broadcasting, land mobile radio, cellular telephones, radar, satellite communications,

remote sensing, and radio astronomy all depend upon the shared use of the radio spectrum which

has benefitted mankind for the past century. The rules and regulations that enable this sharing are

based on fundamental natural laws (of physics), agreements among proximate users, international

treaties, and domestic public law. Spectrum management and frequency assignment represent the

disciplines and processes used to allocate bands of the spectrum to various radio services and

assign frequencies, each with an associated bandwidth, to individual users. In many cases, users

are expected to ensure that their transmitter’s emissions do not adversely affect existing users.

This approach is based on the fact that all electromagnetic signals can be both electronically

generated and separated by the frequency of those signals. Essentially, all conventional signals

use a single-frequency signal – a sinusoidal (sine) wave called a carrier that is modulated with the

information it is to “carry.” Its amplitude, frequency, or phase is varied according to the

information (e.g., voice, video, or data) to be carried.

Radio and wireless communications are managed using tools and techniques that describe signals

in the frequency domain; however, it is very important to recognize that every signal exists

simultaneously in both the frequency domain and the time domain. These domains are simply

alternative ways of describing and processing electronic and electromagnetic (radio) signals. This

is easy to visualize. The cycling of a simple sine wave is the time domain perspective and its

existence at a single frequency is the frequency domain perspective.

Using a carrier allows good control over the bandwidth any signal occupies and has been an

enormously effective approach for dividing the radio spectrum by bands and channels, which has

enabled tractable and effective sharing of the spectrum. It has always been possible, however, to

generate signals without a carrier. In the case of UWB, these signals are simply pulses of

electromagnetic energy shaped by electronic circuitry and a transmitting antenna. Recent

advances in electronics and microcircuits have allowed the development of communications and

radar systems that use such carrierless pulses. It is a fundamental physical law that the narrower

the pulse in the time domain, the wider the emission in the frequency domain.

A classic radar is an example of a device that requires signal processing in both the frequency

domain and the time domain. It has a carrier. Pulses of that carrier are transmitted periodically.

Reflections of that signal from a target return to the point of origin. The radar receiver uses a

filter in the frequency domain to select only that small portion of the spectrum where the radar

signal exists. Then, the receiver uses time domain processing to determine how long it took the

reflection to return and thus determine the distance to the target.
1-3

1.1 Objectives of this Work

The primary objective of this investigation was to develop an understanding of UWB signal

characteristics based on several currently available devices. Both temporal and spectral

characteristics of the UWB signals were sought, with the latter being of particular interest. The

nature of the emission spectrum, whether smooth, comprised of lines, or a combination of both,

is needed for interference analyses. How that emission spectrum depends on the temporal

characteristics such as UWB pulse width, type of signal modulation, and the use of dithering may

be needed to develop sharing policies and regulations. Finally, the nature of the aggregate of

many individual UWB signals is important in understanding how the radio spectrum might be

affected if and when large numbers of UWB devices are deployed.

Practical and repeatable measurement methods to obtain values for particularly useful UWB

signal parameters with available commercial-off-the-shelf (COTS) test equipment may also be

needed for compliance testing related to regulation. Where possible, these practical methods are

identified and any limitations are described. Highly accurate time domain measurements are used

to ensure that the COTS-based measurement methods are reliable.

Finally, an initial assessment of the effects of UWB signals on existing systems will indicate if,

and how much, additional work is needed in this area. A limited effort to determine how much

UWB signal power can pass through the front-end (antenna, amplifiers, and filters) of selected

receivers provides a basis for more detailed investigations of the effects on victim system

performance and allows the calculation of desired signal-to-noise and interference-to-noise ratios

for the selected systems.

1.2 Specific Ultrawideband Systems Measured 

The actual UWB emitters used in this work were borrowed from a number of sources, including

UWB device manufacturers and owners of systems that contain UWB devices or that use UWB

signals to perform their functions. These included prototype, experimental, and operational

systems. Since the objectives of this work were to understand and characterize the radiated

signals and not to evaluate the performance of the systems, the sources of UWB equipment are

not identified. Of the dozen or so devices available, five were selected for the measurements

described in subsequent sections of this report, and are labeled with letters, e.g. Device A. For

comparison, the emissions of an electric drill were also characterized. It is identified simply as

“electric drill.” The devices selected are intended to provide a realistic sample of the various

UWB signal structures being used today.

1.3 Organization of this Report

This investigation of UWB signal structure involved a number of aspects ranging from

theoretical analyses to measurements, both in the laboratory and field, and computer simulations.

As is often the case with broad investigations such as this, a number of workers with different
1-4

skills were involved. To give proper credit to the researchers in each area, the author (or authors)

of each major section of the report is identified at the beginning of each section. The editor was

responsible for assembling the full report.

The first two sections provide orientation and background for the reader. This section contains

some essential background information and the objectives of the work. Section 2 provides the

reader with a brief technical description of UWB technology and some of its salient applications;

this same section also gives a brief history of UWB development and early applications. It also

contains a brief overview of the regulatory issues.

Sections 3 and 4 examine the UWB signal from first principles. Using typical temporal

characteristics of UWB waveforms, the associated spectral characteristics are derived in Section

3. Then in Section 4, the characteristics of a group of individual UWB signals, called the

aggregate signal, is examined.

Sections 5 through 7 describe a variety of measurements of UWB signals and their effects on

selected receivers. Section 5 describes the procedures and results for fundamental measurements

in the time domain. Where possible, the waveform of individual pulses is obtained. Section 6 

describes procedures for, and results of, making similar measurements using commerciallyavailable test equipment. This section also describes procedures for, and results of, band limited

spectral measurements. Section 7 describes the effects that were observed in the receivers of

several Federal Government systems. These measurements do not include an assessment of the

overall performance of those systems; only those effects that are observable in the radiofrequency (RF) or intermediate-frequency (IF) sections of those systems.

Section 8 summarizes the observations made throughout this investigation. These observations

include: the general character of UWB signals (spectra) based on theoretical analyses (Sections 3

and 4); the nature of the actual UWB pulses, both conducted and radiated (Section 5); the nature

of the UWB signal in both the time and frequency domains when received in a range of

bandwidths (Section 6); and the effects on selected receivers (Section 7). Other observations

include which procedures may be best suited for other laboratories that may have only

commercial-off-the-shelf (COTS) test equipment; and the effects various detectors have on

measurements. 

Section 9 contains a comparison of results from measurement, theory, and simulation. 

The Appendices to this report contain supporting information and detailed measurement results.

Appendix A is a brief tutorial on the amplitude probability distribution (APD) which was chosen

as a key measurand for this work. Appendix B describes simulations on the UWB signal

temporal and spectral characteristics of a UWB signal when passed through a limited bandwidth

(receiver or test instrument). Appendix C describes how to convert and/or correct certain

measured values. Appendix D contains the measured data for the five UWB devices and an

electric drill. Appendix E contains the measured data for an aggregate of two, independent UWB

signals.
1The authors are with the Institute for Telecommunication Sciences, National

Telecommunications and Information Administration, U.S. Department of Commerce, Boulder,

CO 80305.

2The term “radio” did not exist until 1912 [2]. It is a shortened form of “radioconductor”

(a contraction of radiation conductor). “Wireless” was the common term before 1912.

2-1

2. UWB TECHNOLOGY AND REGULATORY ISSUES

William A. Kissick and Robert J. Matheson1

The term “ultrawideband” refers to the spectral characteristics of this technology and originates

in the work that led up to a Department of Defense (DoD) study [1]. Alternative terms for the

same technology include impulse radar, impulse radio, carrierless, carrier-free, time-domain, and

others. The fundamental principle is that a short (in time) pulse, also called an impulse, is

generated, transmitted, received, and processed. A fundamental principle, true for any radio

signal, is the relationship between pulse duration and the bandwidth occupied by that signal. 

According to the theoretical Fourier transform, a pulse of duration T seconds (in the time

domain) has an occupied bandwidth of 2/T Hertz (in the frequency domain). For example, a

pulse on the order of a nanosecond in the time domain occupies about two gigahertz of

bandwidth in the frequency domain. An example of time domain signal processing is pulseposition modulation (PPM). Consider the transmission of a train of pulses equally spaced in time.

The receiver processing determines whether each received pulse is located where expected or

arrives early or late. With PPM, a slightly retarded pulse could represent a “0” and a slightly

advanced pulse could represent a “1” when transmitting digital information.

2.1 History of Ultrawideband Technology 

One could say that the first wireless2 system demonstrated by Gugliermo Marconi in 1897 [3],

meets the description of UWB radio. Marconi’s earliest spark-gap transmitters occupied a large

portion of the spectrum, from very low frequencies up through the high-frequency (HF) band and

beyond. And, these systems used manual time domain processing. Morse code was sent and

received by human operators.

The foundations of modern UWB systems were laid down in work done at the Sperry Research

Center in the 1980's by Ross [4]. The emphasis was on the use of UWB as an analytical tool to

explore the properties of microwave networks and to determine the intrinsic properties of

materials [4,5]. These techniques were then logically extended to support experimental analysis

and synthesis of antenna elements [6,7]. These early successes led to the development of an

indoor system to measure the impulse response properties of targets or obstacles [8]. This
3OET Docket 98-153, NOI issued Sept 21, 1998.

2-2

approach of using “short-range radar” obviated the need for an expensive anechoic chamber to

study radar targets, since unwanted reflections from walls and ceilings could be removed by

time-gating techniques.

The use of UWB, with its time domain processing techniques, filled an important need in the

early days of computer development. The appearance of high-speed, sub-nanosecond logic

circuitry in the late 1960s and early 1970s made higher speed computation possible. However, it

was necessary to deliver and distribute large amounts of digital data between the computer

central processor and various input and output devices. This problem was solved by using

multiplexing of multiple signals on a single transmission line using time-domain processing

methods described in a patent by Ross, et al. [9]. This patent could be viewed as a key element in

the foundation of UWB communications. It is a small step from this work to developing wireless

UWB communications. Further developments during the 1970s led to a more thorough

development of principles needed to fully describe and develop the field of time-domain

electromagnetics [10, 11, 12].

In the 1980s and 1990s the principles of time domain electromagnetics were applied to wireless

communications, in particular to short-range communications in dense multipath environments.

Schotz [13] describes this application in detail and explores the advantages and disadvantages.

He showed that a large number of such systems could operate in the same space and that such

wide bandwidth signals are more immune to the deleterious effects of multipath than are narrow

bandwidth signals. A potential application for UWB communications is the accommodation of

many users in high-multipath environments, but the challenge is coexistence within the already

highly-populated radio spectrum. The advantages may or may not outweigh the disadvantages,

and other approaches to wireless operation in dense, high-multipath environments may perform

as well as the UWB approach.

The other major application area of UWB technology is sensing, with the likely niche being

short-range, high-resolution radar. This area requires much less signal processing and uses much

simpler electronics, but has not received as much attention as the more complex communications

applications. Ground penetrating radar was one of the first applications [14]. In 1974, Morey [15]

patented a radar system that, due to the use of a very wide band of frequencies, was able to

penetrate the ground to distances of one to several meters. This patent was later the basis of a

commercial success.

2.2. Regulatory Issues

After receiving three requests by UWB developers, the Federal Communication Commission

(FCC) issued a Notice of Inquiry (NOI)3, to gather information on the possible uses of UWB

devices. Many comments were received in response to that NOI. The FCC also issued the
4Three waivers granted by Chief of OET, June 25, 1999, after coordination with NTIA.

2-3

requested three waivers for a limited number of each of the three low power UWB devices after

coordination on the technical limitations required by NTIA to approve the proposals4.

Information gathered by that NOI led the FCC to release a Notice of Proposed Rule Making

(NPRM) in May 2000. The major regulatory issues in that NPRM are centered on the question of

how much interference UWB systems might cause to existing radio systems. 

The FCC and NTIA jointly manage the radio spectrum in the United States. Part 15 of Volume

47 of the Code of Federal Regulations (47 CFR- Part 15) contains the FCC rules for authorizing

non-licensed operation of low power radio devices that typically radiate signals in bands licensed

for other types of devices. The current Part 15 rules define three classes of radiators: Incidental

Radiators (which do not deliberately generate the RF signals they emit and are not regulated; e.g.,

an electric drill), Unintentional Radiators (which need to generate RF signals, but do not intend

to radiate them, e.g., a computer), and Intentional Radiators (which deliberately radiate low-level

radio signals, e.g., a garage door opener). The NPRM proposes that UWB devices be operated

under a new section of the Part 15 rules, with approximately the same numerical limits for new

UWB devices as for the existing intentional radiators.

Major regulatory issues include a determination of what numeric limits should apply to UWB

emissions and what techniques should be used to measure those emissions. The NPRM proposes

numerical limits and measurement techniques identical to those described in current Part 15 rules

for Intentional Radiators, with the addition of a maximum total absolute peak limit or a possible

peak limit measured in a 50-MHz bandwidth.

Important related questions include whether these limits should be lower in specific restricted

frequency bands used by the Federal Government for particularly critical applications, including

the Global Positioning System (GPS). These critical frequency bands have already been

identified in the existing Part 15 rules, and Intentional Radiators are prohibited from deliberately

radiating signals in any of these identified critical bands. Since UWB systems will typically

radiate energy in frequency bands managed by NTIA, as well as frequency bands managed by the

FCC, the two agencies must concur on the new rules.

2.3 References

[1] “Assessment of Ultra-Wideband (UWB) Technology,” July 13, 1990. DTIC No.

ADB146160. The Executive Summary of this report is published in the IEEE Aerospace and

Electronics Systems Magazine, Nov. 1990. pp 45-49.

[2] W.F. Snyder and C.L. Bragaw, “Achievement in Radio,” NBS Sp. Pub. 555, Oct. 1986.

[3] W.L. Weeks, Antenna Engineering, McGraw-Hill, 1968.
2-4

[4] G.F. Ross, “The transient analysis of certain TEM mode four-post networks,” IEEE Trans. on

Microwave Theory and Tech., Vol. MTT-14, p. 528, Nov. 1986.

[5] A.M. Nicholson and G.F. Ross, “The measurement of the intrinsic properties of materials by

time-domain techniques,”IEEE Trans. on Instrum. and Meas., Vol. IM-9, pp. 377-382, Nov.

1970.

[6] G.F. Ross, “A new wideband antenna receiving element,” 1967 NREM Conference

Symposium Record, Boston, MA, Nov. 1967.

[7] P. Fenster and G.F. Ross, “Members of the class of TEM-mode wire antennas with TLIR”

presented at the IEEE International 1968 PGAP Symp., Boston, MA, Sept. 1968.

[8] A.M. Nicholson and G.F. Ross, “A new radar concept for short-range application.”

Proceedings of IEEE first Int. Radar Conf., pp. 146-151, Washington, D.C. Apr. 1975.

[9] G.F. Ross, L. Susman, and E. Eves, “Time-domain multiplexer-demultiplexer for digital

transmission of GHz data,” U.S. Patent No. 3,763,318, Oct. 1973.

[10] G.F. Ross, “A time-domain electromagnetics bibliography,” Sperry Research Center, Report

SCC-RR-75-1, Mar. 1975.

[11] C.L. Bennett and G.F. Ross, “Time-domain electromagnetics and its applications,”

Proceedings of IEEE, Vol. 66, No. 3, Mar. 1978.

[12] E.K. Miller (Ed), Time domain measurements in electromagnetics, Van Nostrand Reinhold,

1986.

[13] R.A. Scholtz, “Impulse Radio,” IEEE PIMRC 97, Helsinki, Finland, 1997.

[14] D.L. Moffat and R.J. Puskar, “A subsurface electromagnetic pulse radar,” Geophysics, Vol.

41, pp. 506-518, 1976.

[15] R.N. Morey, “Geophysical survey system employing electromagnetic impulses,” U.S. Patent

No. 3,806,795, Apr. 1974.
1The author is with the Institute for Telecommunication Sciences, National

Telecommunications and Information Administration, U.S. Department of Commerce, Boulder,

CO 80305. 

3-1

(3.1)

3. ANALYTICAL DESCRIPTION OF TIME AND SPECTRAL CHARACTERISTICS

OF ULTRAWIDEBAND SIGNALS

Roger A. Dalke1

3.1 Introduction

A theoretical analysis of UWB signals can provide important insights into how UWB emissions

affect various types of RF communications devices. In addition to allowing for direct calculation

of interference effects, analytical results can be used to aid in the planning, design, and validation

of measurements. This section details the results obtained from an analysis of proposed UWB

pulse position modulation schemes. 

The approach used in the analysis and the results are presented in this section. The mathematical

details will be published elsewhere.

3.2 Power Spectrum of UWB Signals

The power spectral density is the average power in the signal per unit bandwidth and hence

provides important information on the distribution of power over the RF spectrum. The power

spectral density for a UWB pulse position modulation scheme using short duration pulses

transmitted at some nominal pulse repetition rate (PRR) is given in this section. The pulse

position is randomized or dithered with respect to the nominal pulse period. The randomization

scheme analyzed in this section is referred to as fixed time-base dither.

3.2.1 UWB Signals Using Fixed Time-base Dither

In the fixed time-base dither scheme, each pulse occurs at the nominal pulse period, , minus a

time increment randomly distributed over a fraction of the nominal period as given in Equation

3.1. This expression also includes binary pulse modulation as proposed for communications

applications.
3-2

(3.2)

(3.3)

(3.4)

where represents the pulse shape that corresponds to an information bit (e.g., represents

the value 0, represents the value 1). The coefficients are related to whether the nth

information bit has the value 0 or 1 as follows:

where are the information bit probabilities (i.e., is the probability of a bit having the value

0, and is the probability of a bit having the value 1). Finally, the random variables 

define the pulse randomization or dithering and are described by a density function , where

For fixed time-base dither, the random variables and are each assumed to be independent

and identically distributed (iid).

It should be noted that the signal given in Equation 3.1 is quite general in terms of the pulse

shape, binary modulation method, and pulse randomization statistics. Hence, the results

presented in this section can be used to predict the power spectral density at various points in the

radio link between an interfering UWB transmitter and a victim receiver (e.g., at the output of

the UWB transmitter, the UWB signal radiated from a particular antenna, or in the IF section of a

narrowband RF receiver). When dealing with linear systems, the various pulse shapes are simply

related by convolutions with the appropriate transfer functions.

The power spectral density is the Fourier transform of the autocorrelation function. The

autocorrelation function is obtained by taking the expected value of the signal at two different

times which is expressed mathematically as 
3-3

(3.5)

(3.6)

Taking the expectation in Equation 3.4 yields

were the symbol is the convolution operator and is the time lag. Functions given in

upper case letters ( ) are the Fourier transforms of the pulse and dithering functions.

The statistics for this process are periodic with period as is evidenced by Equation 3.5. Such

processes are commonly referred to as cyclostationary. Essentially this means that the statistics

depend upon when the process is observed during a period. The victim receiver may observe the

process at an arbitrary time during a period and hence it is useful (and simplifying) to calculate

the average over all possible observation times within a period. Taking the time average over one

period and the Fourier transform of Equation 3.5 yields the average power spectral density of the

fixed time-base dithered UWB signal

The power spectral density has both discrete and continuous components that depend on the

pulse spectrum and the Fourier transform of the density function used to randomize the signal.

Note that when is small at multiples of the PRR, the discrete components are small and the

spectrum is predominantly continuous. When approaches one (negligible dithering) and
3-4

(3.7)

 the bits do not change (e.g., ), the continuous spectrum disappears, and the line spectrum

dominates. The quantity is the expected value of the pulses. 

If bit values are equiprobable (i.e., ) and the pulse representing a 1 is a time delayed

version of the pulse representing a 0 (i.e., ), equation 3.6 reduces to

When the information bit time delay is small relative to the to the dithering delay (i.e.,

 over the range of frequencies for which is significant), the effects of pulse

position modulation on the power spectrum are inconsequential.

The results of an example calculation using Equation 3.7 are shown in the following figures. For

this example, the signal consists of a short-duration pulse (Figure 3.1) transmitted at a 10 MHz

PRR. The dithered pulse position is random and uniformly distributed over 50% of the pulse

period. In this calculation, it is assumed that the effects of information bit modulation are

negligible over the frequency range of interest. The power spectral density over a frequency range

of 1-5000 MHz is shown in Figure 3.2. The magnitude of the spectrum is normalized to the peak

of the continuous distribution (at about 250 MHz). The Fourier transform of the density function

for this example is . This function has nulls at frequencies equal to 

( ), hence the interval between discrete spectral lines is 20 MHz as shown in the

figures. For frequencies above 20 MHz, the continuous spectrum is approximately the same as

the pulse spectrum (i.e., ). Figure 3.3 shows the discrete spectrum over a more limited

range (800-1600 MHz) to highlight the individual spectral lines.
3-5

-0.25

0

0.25

0.50

0.75

1.00

0 1 2 3 4 5

Time (ns)

Voltage

Figure 3.1. Time domain pulse shape.

-40

-20

0

20

40

60

80

0 1000 2000 3000 4000 5000

Discrete spectrum

Continuous spectrum

Frequency (MHz)

dB

Figure 3.2. Power spectral density for a fixed time-base dithered 10 MHz

UWB signal. The pulse positions are uniformly distributed over

50% of the pulse repetition period.
3-6

-40

-20

0

20

40

60

80

800 1000 1200 1400 1600

Discrete spectrum

Continuous spectrum

Frequency (MHz)

dB

Figure 3.3. Power spectral density showing discrete and continuous spectrum

from 800 to 1600 MHz.

The mean power in the bandwidth of a narrowband victim RF receiver as a function of frequency

can easily be calculated from these results. For example, Figure 3.4 shows the power available to

a receiver with a nominal 10 kHz bandwidth. As shown in the figure, the discrete spectrum is not

a factor for RF frequencies above a few hundred MHz. For narrowband victim receivers where

gains due to the UWB transmitter filters/antenna, propagation channel, and receiver are fairly

constant over the receiver bandwidth, the received interference power can easily be calculated by

applying the appropriate gain factors to the power in the receiver bandwidth at the center

frequency of the receiver.
3-7

-40

-20

0

20

40

60

80

0 500 1000 1500 2000

Power in 10 kHz

Discrete spectrum

Continuous spectrum

200 MHz

Frequency (MHz)

dB

Figure 3.4 Power spectral density showing the continuous spectrum in a 10

kHz bandwidth compared to the discrete spectrum.

(3.8)

3.2.2 Power Spectrum for Finite Duration and Repeated Signals

The results based on Equation 3.1 assume that the signal is on continuously. Obviously, real

signals are of finite duration. Also, for some proposed systems, the signal is transmitted for a

length of time, say , and then repeated. In this section, we extend the results presented above

to finite duration and repeated signals.

To obtain the power spectrum for a finite duration signal, the following window function 

is multiplied by (Equation 3.1). The result is that given in Equation 3.7 convolved with the

spectrum of the window, i.e., as may be expected. As the window duration

increases, the spectrum shape approaches .

When the series is windowed and repeated, the autocorrelation function is obtained by

taking the expectation of periodic extension of a windowed portion of the series or
3-8

(3.9)

(3.10)

(3.11)

The resulting spectrum is 

which is now discrete with spectral lines at frequency intervals of .

3.3 Band Limited Signal Statistics for Fixed Time-base Dithered Systems

From the standpoint of a victim receiver, a fixed time-base dithered UWB signal is a random

process. A knowledge of the statistics of such a process is important in predicting how

interference affects the performance of a victim receiver. When the UWB PRR is larger than the

receiver bandwidth, it may be expected that the received signal would appear to be

indistinguishable from Gaussian noise. Since receiver performance in a Gaussian noise

environment is well understood, quantifying conditions for which the received UWB interference

resembles Gaussian noise is important in predicting receiver performance and developing

emissions requirements. Also, when the received signal is Gaussian, only one parameter (mean

power) is required to characterize the process. In this section we present the results of an analysis

of the fixed time-base dither scheme that can be used to predict when the received UWB signal is

approximately Gaussian.

For this analysis, we seek to determine the probability density function that describes the

statistics of the UWB signal as seen by the victim receiver (e.g., the final IF stage of the

receiver). The following relationship between the density function , its characteristic

function , and the pulse randomization density function is used to obtain an

approximate expression for the received signal statistics

Formally, the desired density function is obtained by inserting the UWB signal (Equation 3.1)

into Equation 3.11 and taking the inverse Fourier transform of the characteristic function. 
3-9

(3.12)

(3.13)

(3.14)

The characteristic function is periodic since the process is cyclostationary as discussed in Section

3.2.1. For purposes of this analysis, the time averaged statistics are obtained by averaging over a

period as with the power spectral density function 

After some manipulations, the density function can be expanded into the well known Edgeworth

[1] series. The first four terms of the series are 

where

The desired density function is related to by using the transformation

where is the mean and is the standard deviation, hence .

The first term in the series is the standard normal distribution. The following terms are scaled by

coefficients known as the skewness and excess [1 ]. 

In general, the skewness and excess are rather complicated functionals of the pulse shape and

the pulse randomization statistics . In the case of a narrowband receiver with a center frequency

larger than twice the PRR, the expressions are greatly simplified. The following results assume

that the power in the spectral lines (if present) is much smaller than that due to the power in the

receiver bandwidth due to the continuous spectrum. In addition, if the UWB pulse 

spectrum is approximately constant over the bandwidth of the receiver, the variance ,

skewness, and excess can be expressed in terms of the baseband impulse response of the receiver

filter, , as follows:
3-10

(3.15)

-2

0

2

4

6

10

1

10

2

10

3

10

4

10

5

10 MHz

Bandwidth (kHz)

Excess

Figure 3.5. The excess as a function of receiver bandwidth.

These results show that the variance is proportional to the receiver bandwidth as expected. The

mean and skewness are negligible due to the oscillatory characteristics of the bandpass filtered

signal. The behavior of the excess as a function of receiver bandwidth was calculated for a

receiver with a raised cosine lowpass characteristic and a UWB signal with a 10 MHz PRR. The

signal is dithered uniformly over 50% of the pulse repetition period. 

Figure 3.5 shows the excess as a funtion of receiver bandwidth. Note that the distribution is

approximately Gaussian up to about a 1MHz bandwidth. The excess then decreases to a

minimum at about 20 MHz, after which it increases. The normalized distribution for bandwidths

below 1 MHz and at10 and 20 MHz are shown in Figure 3.6. 
3-11

0

0.1

0.2

0.3

0.4

0.5

-4 -2 0 2 4

γ

2

≈ 0 (f < 1 MHz)

γ

2

= -1.26 (20 MHz)

γ

2

= -.938 (10 MHz)

Figure 3.6. The distributions for various receiver bandwidths of less than

1 MHz, and bandwidths of 10 and 20 MHz. 

The results presented in this section can be used to predict when an interfering fixed time-base

dithered UWB signal is approximately Gaussian in nature, and hence, should be useful in

providing guidance to system designers and regulators. Furthermore, as shown in the previous

example, they can be used to estimate statistics for bandwidths comparable and exceeding the

UWB PRR. In cases where the bandwidth is much larger than the PRR, so that the receiver

actually resolves the individual pulses, the results presented above are no longer valid. In such

cases, amplitude statistics can readily be estimated by calculating the fraction of time that a

particular pulse (as seen by the receiver) amplitude is exceeded during the pulse repetition

period.

3.4 References

[1] Harald Cramer, Mathematical Methods of Statistics, Princeton NJ: Princeton University Press,

1945.
This Page Intentionally Left Blank

This Page Intentionally Left Blank
1The author is with the Institute for Telecommunication Sciences, National

Telecommunications and Information Administration, U.S. Department of Commerce, Boulder,

CO 80305. 

4-1

(4.1)

4. CHARACTERISTICS OF AN AGGREGATE OF ULTRAWIDEBAND SIGNALS

Roger A. Dalke1

4.1 Introduction

The proliferation of UWB devices throughout the United States has been predicted by many

industry sources. Hence, it is important that the effects of an aggregate of such devices on RF

spectrum users be well understood by regulators, spectrum users, and UWB system designers. 

This section describes models that can be used to predict interference effects of many UWB

devices on traditional narrowband RF receivers.

This model assumes that the victim receiver is narrowband and hence, it is sufficient to evaluate

UWB parameters such as effective isotropically radiated average power (EIRP) and antenna

gains at the center frequency of the receiver. The calculation of the power at the victim receiver

requires an estimation of the basic transmission loss over the propagation path from the

transmitters to the receiver. Single frequency propagation models used in traditional radio link

calculations will be utilized in conjunction with the models described in this section. In the

analysis which follows, it was convenient to use the basic transmission gain (denoted below as

) instead of loss. The basic transmission gain and loss are reciprocals and have the same

absolute value but opposite signs when given in decibels.

In the first part of this section, the aggregate effects of a few similar devices in the immediate

vicinity of a victim receiver are discussed. This is followed by the development of a statistical

model that can be used to calculate the average received power from many UWB devices

randomly distributed over the surface of the Earth. This model can be used to predict interference

power for both terrestrial and airborne receivers.

4.2 Deterministic Interference Model for UWB Devices in the Vicinity of a Victim Receiver

The mean power in the receiver bandwidth due to UWB devices is simply the sum of the power

received from each source or 
4-2

(4.2)

where is the received power, is the emitted EIRP in the receiver bandwidth, is the

transmitter gain, and is the receiver gain in the direction of the nth transmitting device. When

the locations of the devices are known and N is small, computing the received power is a

relatively straightforward matter. 

More realistically, one may have only a rough estimate of the ostensible number of such devices

deployed in a particular geographic area (e.g., an average areal density) surrounding a particular

RF receiver. In such cases, Equation 4.1 is not very useful since the parameters (perhaps the most

important one being ) are not known. Hence statistical models and estimates are required to

make any progress in predicting the potential for interference. The development of such a model

is given in the next section. 

Equation 4.1 is valid for commonly encountered random RF signals because the variance of the

sum of zero mean random variables is the sum of the individual variances. When the received

signals are normally distributed, the mean power is all that is needed to describe the statistics of

the resulting interference. If there are many such devices with the same statistical properties (not

necessarily normally distributed) then the statistics of the sum will approach a normal

distribution [1]. In such cases, the models that predict the mean interference power provide the

only statistic necessary to describe the process.

This leads directly to the question of how many signals must be added before the aggregate signal

realistically appears to be normally distributed. Perhaps some insight can be gained by examining

the results for a band limited fixed time-base dithered UWB signal as described in Section 3.3. In

this example, the signal statistics are approximately normal for bandwidths well below the PRR.

As the bandwidth increases, the absolute value of the excess increases and the statistics are no

longer normal. The excess for an aggregate of such devices can be calculated as described below.

The aggregate excess for the sum of several random variables is related to the excess of each

random variable as follows

where is the second central moment of each variable and is the sum of the moments.

Since the processes are zero mean, the second central moment is just the signal power given in

Equation 4.1.
4-3

(4.3)

(4.4)

(4.5)

The aggregate excess for band limited signals (e.g., as given in Section 3.3) is therefore

When the individual excesses, EIRP, and gains are the same, Equation 4.3 reduces to the well

known result

which indicates that the excess can decrease fairly rapidly as additional devices are added.

4.3 Statistical Aggregate Model

In this subsection, we develop a statistical model that can be used to estimate received

interference power from many devices randomly distributed over the area surrounding a victim

receiver. It is assumed that the devices are uniformly distributed over the surface of the Earth.

The model requires an estimate of the path gain over the geographical area surrounding the

receiver, the average receiver and transmitter antenna gains, and the average areal density of

transmitters. The areal path gain can be calculated from traditional propagation models such as

the Irregular Terrain Model [2]. A simple methodology that can be used to estimate average

transmitter antenna gain is given in this subsection. Example calculations are given using simple

receiving and transmitting antennas.

Let a UWB device with EIRP and gain be located at a point in space denoted by . The

gain due to free space and terrestrial propagation from the point to the victim receiver, , is a

random variable that depends on location, terrain, climate, and other factors. Assuming a receiver

gain , the received power is

where represents the dependence on the spatial location and are points in some 

probability space that characterizes, for example, random variations in devices, how, when and

where they are deployed, propagation paths, etc. The average power at the victim receiver due to
4-4

(4.6)

(4.7)

(4.8)

(4.9)

many UWB devices is obtained by taking the expected value of the sum of the contribution from

each device

where is an area increment at the point and is the number of devices in .

In this model, it will be assumed that the devices are randomly distributed in space according to

Poisson Postulates. Essentially this means that the number of devices in non-overlapping regions

of space are independent, the probability structure is both space and time invariant, and the

probability of exactly one device being in a small increment of space is approximately

proportional to the increment

where is the average density. The probability of more than one device being in a small interval

is smaller than the order of magnitude of (i.e., ). The average received power is then

The expected values of the transmitted power and gain will depend, for example, on the

range of possible devices and the antenna orientations with respect to the victim receiver. The

mean path gain is a function of the space coordinates. The mean receiver gain will also in

general be a function of the space coordinates. Assuming a distribution in 2-space corresponding

to the surface of the earth, for small increments, the received power can be calculated via

integration. Using polar coordinates with the victim receiver located at the origin, the average

power (assuming and are independent) is 
4-5

(4.10)

(4.11)

In this expression, the basic path gain is the average over all possible radial paths and may be

calculated, for example, by using the Irregular Terrain Model in the area prediction mode. The

integral over includes the directive gain of a typical receiver. In this model, the parameter is

constant and is equal to the average number of devices per unit area.

4.3.1 Example Calculation Using the Irregular Terrain Model (ITM)

Converting Equation 4.9 to decibels, we have

As is customary, upper case letters are used to denote decibel equivalents . The mean transmitter

power can be estimated from specifications or measurements of typical UWB devices. 

A Method for Estimating 

In this model, it is assumed that the transmitting antennas are randomly oriented. The average is

obtained by assuming a probability distribution for the orientations and applying a typical UWB

transmitter gain function which can be defined in terms of the usual spherical coordinate system

angles and . In what follows, is the angle from the pole of the sphere located, for example,

at the top of the transmitter antenna (e.g., the top of a vertical dipole) and is the azimuth.

In the case of a victim receiver near the ground, it is reasonable to assume that the direction of

propagation to the receiver is uniformly distributed over a solid angle defined by a band on

the unit sphere bounded by spherical angles and ( ). The expected value of

the gain in the direction of the victim receiver in terms of the directive gain function is 
4-6

(4.12)

(4.13)

(4.14)

where

The expected value of is then 

As an example, consider a short dipole where . The expected value of the gain

is 

When the transmitters are oriented so that , dB and when , dB.

Calculation of Areal Gain Using ITM

The ITM in area prediction mode was used to obtain the average path gain relative to free

space as function of distance from the victim receiver. The basic path gain was

then integrated to obtain . Table 4.1 gives typical ITM parameter settings used for examples

given below unless otherwise specified.

Referring to Equation 4.10, the basic path gain is integrated over the interval . The usual

free space gain formula is only valid in the far field and has a singularity at . In the near

field (less than a few wavelengths), power is transferred between the antennas via mutual

coupling. For the purposes of this analysis, close proximity free space coupling was

approximated by fitting a function to data obtained from a numerical analysis of the maximum

coupling between two half-wave dipoles in the near field [3]. The resulting function used to

calculate free space gain is
4-7

-40

-30

-20

-10

0

0 1 2 3

Gfs

=-20log(4πx+1.64)

Numerical Analysis

Separation (wavelengths)

Max Coupling - Antenna Gain (dB)

Figure 4.1 Approximation for near field antenna coupling.

(4.15)

which closely approximates near-field results and gives the usual far-field behavior when the

antennas are separated by more than a few wavelengths. The numerical results for near-field

coupling and (maximum coupling less the gain of the half-wave dipoles) as a function of

antenna separation are shown in Figure 4.1.

For large distances, the integration is truncated well into the diffraction region (beyond the

smooth earth radio horizon) where contributions are negligible. Figure 4.2 shows the basic

transmission gain and path gain , obtained from ITM for the parameters given in Table

 4.1.
4-8

-200

-150

-100

-50

0

0 25 50 75 100 125

Gb

Gfs

Gp

Distance (km)

(dB)

Figure 4.2. Example calculation of basic transmission gain using ITM. is

the path gain and is the free-space gain. (1000 MHz, =90 m,

Tx = 2 m, Rx = 3 m).

Effects of and Receiver Height

In area prediction mode, the statistical parameter is used to characterize terrain in the

geographical region of interest. The dependence of the parameter on is shown in Figure

4.3. Of note is the fact that in flat terrain is more than 20 dB greater than for hilly terrain

( m).

In Figure 4.4, the parameter is plotted as a function of receiver height. Basically, the path gain

increases with increasing antenna height since terrestrial attenuation is not a factor at increasing

distances from the receiver (as the receiver height increases). With increasing height, the path

gain from the entire region within line-of-sight of the receiver is essentially due to free space

propagation.
4-9

-80

-60

-40

-20

0

0 50 100 150 200

5 GHz

3 GHz

2 GHz

1 GHz

0.5 GHz

0.1 GHz

Δh (m)

Γ

b

(dB

m

2

)

Figure 4.3. Basic areal gain as a function of and frequency. (Tx = 2 m, 

Rx = 3 m).

-40

-35

-30

-25

-20

-15

0 200 400 600 800 1000

Receiver Height (m)

Γ

b

(dB

m

2

)

Figure 4.4. Basic areal gain as a function of receiver height. (1000 MHz,

=90 m).
4-10

(4.16)

Table 4.1 Parameters for ITM Calculations

ITM Parameter Value

Frequency Various

Receiver Antenna Height 3 m

Transmitter Antenna Height 2 m

Polarization Vertical

Terrain Irregularity Parameter 90 m, 30 m, 0 

Ground Electrical Constants .005 S/m, 

Surface Refractivity 301 N-units

Climate Continental Temperate

Siting Criteria Random

Time and Location Variability 50%

Confidence 50%

Estimated Interference Power Levels for a Half-wave Dipole Receiver

Referring to Equation 4.9, azimuthal dependence of the receiver gain in the direction of the UWB

transmitters can be explicitly included in the analysis. The quantity defined in Equation 4.10 is

just the average gain in the azimuthal direction. To give a simple example, a half-wave dipole has

a constant azimuthal gain of 2.15 dBi, hence .

Assuming that the UWB transmitters are short dipoles and , the power at the receiver per

watt of transmitted power is

where is the average density in dB per unit area, and is area average of path gain. Calculated

valuesfor variousfrequencies and terrain parameters associated with so called flat ( ), plains

( ), and hills ( ) environments are given in Table 4.2 below.
4-11

Table 4.2 as a Function of and Frequency (Based on Parameters Given in Table 4.1)

Frequency (MHz)

100 0.14 -2.51 -11.61

500 -11.46 -18.56 -31.97

1000 -16.84 -27.48 -39.11

1500 -20.03 -32.02 -43.35

2000 -22.32 -34.88 -46.53

2500 -24.10 -36.93 -49.13

3000 -25.56 -38.53 -51.26

3500 -26.81 -39.84 -53.11

4000 -27.89 -40.97 -54.74

4500 -28.83 -41.96 -56.24

5000 -29.69 -42.77 -57.51

4.3.2 Example Calculation Assuming Free Space Propagation to the Radio Horizon

When the victim receiver is located high above the earth, as with an aircraft receiver, the

transmission path to the radio horizon is largely unaffected by the earth (see Figure 4.4). In such

cases, the interfering signal power can be estimated by assuming free space propagation to all

devices located within the radio horizon. It should be noted that the methodology described below

neglects the effects of line-of-sight propagation in the troposphere and that due to diffraction and

tropospheric scatter from beyond the radio horizon. The over-the-horizon diffracted and scattered

signals will be minimal in most cases of interest.
4-12

(4.17)

(4.18)

(4.19)

(4.20)

(4.21)

The areal gain is calculated from

where is the height of the receiver, is the height of the transmitter, and is the distance

to the radio horizon which can be calculated using the following approximate expression [2]

The earth’s effective curvature isthe reciprocal of the earth’s effective radius and is normally

determined from the surface refractivity using the empirical formula [2]

where is the effective earth radius factor, is the surface refractivity, = 179.3 N-units, and

= = 157 × 10!9 m!1 = 157 N-units/km [2].

Evaluating the integral in Equation 4.17 gives

When the receiver height is much greater than the transmitter height the result can be reduced to

Assuming a standard four thirds earth ( ), Figure 4.5 shows the areal gain as a function of

frequency and receiver antenna height as compared with calculated using the ITM up to its

recommended limit of 1 km. Note that at 1 km, the results are within about 0.5 dB. Using Equation

4.21, Figure 4.6 shows for various frequencies as a function of receiver height up to 10 km.
4-13

-40

-30

-20

-10

0

10

0 2 4 6 8 10

5000 MHz

4000 MHz

3000 MHz

2000 MHz

1000 MHz

100 MHz

Receiver Height (km)

Γ

b

(dB

m

2

)

Figure 4.6. Basic areal gain as a function of receiver height assuming freespace gain to the radio horizon.

-40

-35

-30

-25

-20

-15

1 10 100 1000 10000

Free space to horizon

ITM

Receiver Height (m)

Γ

b

(dB

m

2

)

Figure 4.5. Basic areal gain as a function of receiver height using ITM

compared with a direct calculation assuming free-space propagation.

(1000 MHz, =90 m).
4-14

4.4 References

[1] Harald Cramer, Mathematical Methods of Statistics, Princeton NJ: Princeton University Press,

1945. 

[2] G.A. Hufford, A.G. Longely, and W. Kissick “A guide to the use of the ITS irregular terrain

model in the area prediction mode,” NTIA Report 82-100, April 1982.

[3] Constantine A. Balains, Antenna Theory Analysis and Design, 2

nd Edition, New York: John

Wiley and Sons, 1997, p 442.
1The author is with the Institute for Telecommunication Sciences, National

Telecommunications and Information Administration, U.S. Department of Commerce, Boulder,

CO 80305. 

2The authors are with the Radio-Frequency Technology Division, National Institute for

Standards and Technology, U.S. Department of Commerce, Boulder, CO 80305. 

5-1

5. FULL-BANDWIDTH REFERENCE MEASUREMENTS OF ULTRAWIDEBAND

EMISSIONS

Brent Bedford,1 Robert T. Johnk,2 and David R. Novotny2

5.1 Introduction

Ultrawideband (UWB) signals, by definition, contain energy over a larger range of the frequency

spectrum than do conventional radio signals which are relatively narrow-banded. The majority of

conventional radio test equipment, however, are designed to measure the signals from the

majority of radio systems in use, which constitute mostly narrowband and a few wideband

signaling systems. UWB is a new class of signals that places new demands on measurement

equipment.

There exists a need to characterize this new class of signals across their full emission bandwidth.

This section describes measurements that address that need by capturing the pulse shapes and

inner pulse structure from a selection of UWB devices. The UWB signals in this study are very

narrow pulses of RF energy that are modulated or envelope-shaped in various ways. This study

provides a view of the UWB pulses that cannot be directly measured with common narrower

bandwidth equipment. From this full-bandwidth view of the pulses, comparisons can be made

with measurement results performed with conventional equipment. The measurement results in

this section provide a reference to which other measurement results can be compared, to see how

well the reference set of signal parameters can be predicted from measurements using bandwidthlimited equipment.

5.2 Measuring Instruments and Calculation Methods

The signals emitted from a selection of UWB devices (see section 1.2) were measured by the

National Institute of Standards and Technology (NIST) Radio-Frequency Technology Division to

obtain data that represents the radiated time-domain waveform. The goal of these measurements

was to capture a detailed view of a single pulse. The pulses were measured in two different

environments. The first environment was called "conducted measurements." The second

environment was called "radiated measurements."
5-2

(5.1)

(5.2)

Two different measuring instruments were used in making the full-bandwidth measurements.

The first instrument was a sampling oscilloscope. This instrument was capable of achieving very

high equivalent sample rates when digitizing the input signal. The instrument used in this study

possessed a bandwidth of 20 GHz with the ability to acquire 4,096 samples in a single timedomain record. Due to the nature of how this instrument performs its sampling, it had two

limitations. The repetition rate of the pulses to be measured must be constant and the pulse shape

invariant. While some UWB devices satisfy this requirement, there are devices that do not and

were measured by a second measuring instrument.

The second measuring instrument was a single-event transient digitizer. This instrument had the

advantage of placing fewer restrictions on the pulse parameters that it can measure. The digitizer

possessed a bandwidth of 4.5 GHz with a maximum of 1,024 samples in a single shot. The

instrument was designed to perform high fidelity measurements on a single pulse.

Two quantities were calculated from each measured waveform. The first quantity was "Total

Peak Power." Given that there are i sample points in the time-domain waveform and x is the ith

sample point, total peak power was calculated using equation 5.1 and was the maximum ith value

of the power vector.

The second quantity was "Total Average Power." It was calculated as shown in equation 5.2.

where xi is the ith time-domain sample

)t is the sample interval

pri is the pulse repetition interval

The PRI that was used in the calculation is the shortest time interval between any two pulses.

Effects which could lengthen the PRI such as On-Off-Keying or the quiet time between bursts of

pulses were not considered since measuring these parameters was beyond the scope of this

investigation. Some devices could operate with more than one mode setting. For these devices,

the maximum and minimum PRIs were used in the calculation.
5-3

Attenuator

Trigger Signal

3 dB

Power Splitter

0-18 GHz

Sampling Oscilloscope

UWB

Device

Figure 5.1. Device A, conducted measurement

test setup.

Attenuator

Trigger Signal

3 dB

Power Splitter

0-18 GHz

Single-Event Transient Digitizer

UWB

Device

 Figure 5.2. Device B and D, conducted

measurement test setup.

5.3 Conducted Measurements

Two different test setups were implemented for the conducted measurements. The first is shown

in Figure 5.1. The RF output of the UWB device-under-test was connected using a coaxial

transmission line to an attenuator. The attenuator was used to prevent overloading and damage to

the measurement instrument from too strong a signal level. The signal was then split into two

equal amplitude levels and fed into a trigger port and a signal port on a sampling oscilloscope.

Several pulses were measured to check for pulse shape variations that might induce measurement

errors. This setup was used to perform conducted measurements on device A, which has a

constant pulse repetition frequency.

The second test setup is shown in Figure 5.2. The only difference from Figure 5.1 is the use of a

single-event transient digitizer. This setup was used to perform conducted measurements on

devices B and D due to a non-constant pulse repetition rate in the emissions.

The measured time-domain waveform for device A is shown in Figure 5.3. It exhibits a large

main pulse followed by some damped ringing. The vertical axis represents voltage at the RF

output connector of the UWB device. The corresponding frequency-domain power spectrum,

which was calculated from the time-domain waveform, is shown in Figure 5.4. The vertical axis

represents decibels relative to a milliwatt at the RF output connector of the UWB device. The

caption presents a )f number which is the frequency spacing between the graphed points.
5-4

-4

-2

0

2

4

0 5 10 15 20 25

Time (nanoseconds)

Volts

Figure 5.3. Device A, conducted time-domain waveform.

-80

-60

-40

-20

0

4000 5000 6000 7000

Frequency (MHz)

dB

m

Figure 5.4. Device A, conducted power spectrum, )f = 16.67 MHz.
5-5

The following table summarizes the Total Peak Power and the Total Average Power calculated

for the devices for which conducted measurements were performed.

Table 5.1. Total Peak and Total Average Powers from the Conducted Measurements.

Device Letter Total Peak Power (dBm) Total Average Power (dBm)

A 23.1 -27.8

B 32.0 -4.5

D 17.4 -16.0

The -10 and -20 dB bandwidths were extracted from the frequency-domain power spectrum

graphs. These bandwidths, from the conducted measurements, are summarized below.

Table 5.2. Emission Bandwidth from the Conducted Measurements.

Device Letter -10 dB Bandwidth (MHz) -20 dB Bandwidth (MHz)

A 616.6 799.9

B 479.9 539.9

D 1349 2597

Appendix D contains the complete set of conducted measurement graphs for devices A, B, and

D.

5.4 Radiated Measurements

Four different test setups were implemented for the radiated measurements. All of the test setups

were performed in the NIST anechoic chamber. The first test setup is shown in Figure 5.5. The

UWB device-under-test radiates using its manufacturer supplied antenna into the chamber. A

ridged horn antenna was used in this configuration. The measurement frequency range using this

antenna was 1 GHz to 4 GHz. Two stages of amplification were needed in this configuration to

provide enough signal to drive the measuring instrument. A calibration was performed on the

amplifiers to provide a frequency response correction. The signal was then split into two equal

amplitude levels and fed into a trigger port and a signal port on a single-event transient digitizer.

This setup was used to perform radiated measurements on device C. Figure 5.6 shows the second

test setup that was used. The only difference from the previous test setup is a single stage of

amplification since this UWB device produced a stronger signal. This setup was used to perform

radiated measurements on device D. Figure 5.7 shows the third setup. The only difference from

the previous test setup is the use of a different receiving antenna. The receiving antenna was a

NIST 30 cm TEM horn, which produces minimal waveform distortion. The measurement
5-6

Trigger Signal

3 dB Power Splitter

0-18 GHz

Ridged Horn

Single-Event Transient Digitizer

Amplifier

UWB

Device

 Figure 5.6. Device D, radiated

measurement test setup.

Trigger Signal

3 dB Power Splitter

0-18 GHz

Ridged Horn

Single-Event Transient Digitizer

2-Stage

Amplifier

UWB

Device

Figure 5.5. Device C, radiated measurement

test setup.

frequency range using this antenna was 200 MHz to 4000 MHz. This setup was used to perform

radiated measurements on device E (1500 MHz and 900 MHz modes). Figure 5.8 shows the

fourth test setup. The only difference from the previous test setup is the addition of an attenuator

to trim the measurement system gain down to an optimum level for the measurement. The NIST

30 cm TEM horn was used in this measurement as the receiving antenna. This setup was used to

perform radiated measurements on device E (300 MHz mode) and B.
5-7

Trigger Signal

3 dB Power Splitter

0-18 GHz

Single-Event Transient Digitizer

Amplifier

UWB

Device

Figure 5.7. Device E (1500 MHz and 900

MHz), radiated measurement

test setup.

Trigger Signal

3 dB Power Splitter

0-18 GHz

Single-Event Transient Digitizer

Amplifier

UWB

Device

Attenuator

 Figure 5.8. Device E (300 MHz) and B,

radiated measurement test

setup.

The measured time-domain waveform for device C is shown in Figure 5.9. The vertical axis

represents voltage at the receiving antenna terminals. The separation distance between the

receiving antenna and the transmitting antenna was one meter. The corresponding frequencydomain spectrum is shown in Figure 5.10. The vertical axis represents field strength (decibels

relative to a microvolt per meter), calculated from the time-domain waveform, at the receiving

antenna's location. The caption presents a )f number, which is the frequency spacing between

the graphed points.
5-8

-0.02

-0.01

0

0.01

0.02

5 10 15

Time (nanoseconds)

Volts

Figure 5.9. Device C, radiated time-domain waveform.

40

50

60

70

80

90

1000 2000 3000 4000

Frequency (MHz)

Field Strength (dBuv/m)

Figure 5.10. Device C, radiated peak field strength at 1 m, )f = 20 MHz.

The following table summarizes the Total Peak Power and the Total Average Power calculated

for the devices for which radiated measurements were performed.
5-9

Table 5.3. Total Peak and Total Average Powers from the Radiated Measurements.

Device Letter Total Peak Power (dBm) Total Average Power (dBm)

B (maximum PRI) -3.2 -39.8

B (minimum PRI) -3.2 -33.7

C -21.1 -48.6

D (maximum PRI) -20.5 -51.8

D (minimum PRI) -20.5 -41.8

E (1500 MHz) -7.9 -55.6

E (900 MHz) -3.7 PRI unknown

E (300 MHz) 12.5 PRI unknown

.

The -10 and -20 dB bandwidths were extracted from the frequency-domain radiated spectrum

graphs. Some of the devices had combinations of center frequency and bandwidth such that the

portion of the spectrum of interest exceeded the valid frequency range of the radiated

measurements. The bandwidth could not be determined so these cases are marked with "NA".

The bandwidths, from the radiated measurements, are summarized below.

Table 5.4. Emission Bandwidths from the Radiated Measurements .

Device Letter -10 dB Bandwidth (MHz) -20 dB Bandwidth (MHz)

B 319.9 539.9

C 659.8 1080

D NA NA

E (1500 MHz) 2799 NA

E (900 MHz) 1650 NA

E (300 MHz) NA NA

Appendix D contains the complete set of radiated measurement graphs for devices B, C, D, and

E. It is interesting to compare the conducted and radiated results for devices B and D. For both

devices, the radiated and conducted waveforms are significantly different. While device B's

conducted and radiated spectrums look similar, device D's conducted and radiated spectrums are

significantly different over the 1 GHz to 2 GHz regions.
This Page Intentionally Left Blank

This Page Intentionally Left Blank
1The author is with the Institute for Telecommunication Sciences, National

Telecommunications and Information Administration, U.S. Department of Commerce, Boulder,

CO 80305.

2COTS equipment is defined as measurement devices that are commercially available.

6-1

6. BANDWIDTH LIMITED MEASUREMENTS OF ULTRAWIDEBAND DEVICE

EMISSIONS

Frank Sanders1

6.1 Introduction

In many cases, the characteristics of UWB signals must be measured using equipment that is

bandwidth limited. I.e., the measurement bandwidth is less than the UWB emission bandwidth. 

Bandwidth limited measurements are necessary for at least three reasons:

• Coupling between UWB emissions and various types of radio receiver equipment

will generally be bandwidth limited (by either the receiver RF front-end or the

receiver IF section). Bandwidth limited measurements closely match the case of

bandwidth limited coupling into receivers.

• Regulatory standards may be specified in particular bandwidths. Compliance

measurements must be performed in (or extrapolated to) the required

bandwidths.

• Adequate full-bandwidth measurement systems are not likely to be available to all

measurement facilities. Even if such equipment is available, the outputs

generated may not be satisfactory for all measurement purposes. 

In this section, measurement techniques are described that may be generally applied with

commercial, off-the-shelf (COTS) equipment.2 COTS-compatible methods are included for

measuring the following UWB emission parameters:

• emission spectra as a function of IF measurement bandwidth,

• pulse width estimation,

• pulse shape as a function of IF measurement bandwidth,

• pulse repetition rates, sequences, and gating,

• amplitude probability distributions as a function of IF measurement bandwidth,

• peak power,

• average power.
3Unless otherwise noted, all measurement techniques described in this section have been

implemented by ITS and have been used to generate the data that appear in Section 8 and

Appendix D. 

4Emission width may be defined at the 3 dB, 6 dB, 10 dB, 20 dB points, etc., in the

appropriate domain.

5This discussion of time domain measurements assumes that the UWB RF emission is

rectified by a diode or equivalent detector, either as a discrete component or as part of a system

such as a spectrum analyzer. For other types of time domain measurements, including those that

preserve phase information, see Section 5 of this report.

6-2

This section provides guidance for other laboratories in the implementation of UWB emission

measurement techniques. Each technique is described both generically and as performed

specifically at ITS using COTS equipment.3 Based upon experience gained at the ITS laboratory,

the technical strengths and weaknesses of each approach are described. Particular problems that

other laboratories may encounter with these techniques are noted, along with any practicable

solutions developed by ITS measurement personnel.

6.2 Bandwidth Limited Measurement Theory 

A measurement result is the convolution of an input signal with the impulse response of the

measurement device in the appropriate domain (e.g., time or frequency). The convolution width

of a measurement device’s impulse response may be wider than, equal to, or narrower than the

input signal function. For UWB device emissions, the measurement convolution width is

generally narrower than the device emission.4

 However, there may exist within the UWB emission

some features that are individually narrower than the measurement convolution. These and other

characteristics of UWB emissions must be taken into account in emission measurements, as

described below.

6.2.1 Bandwidth Limited Time Domain Measurement Theory5

The convolution bandwidth of a time domain measurement may be limited by either the RF 

front-end or the time domain digitizer. If the RF front-end is an instrument such as a spectrum

analyzer, then it will be the limiting factor. If the front-end is a wideband detector diode 

(typically a discrete component with 18 GHz bandwidth), then the digitizer will limit. In either

case, the measurement system response will show time domain features that are limited by the

fastest response of the slowest component. That is, a 500-MHz single-shot digitizer (with a

wideband detector) will produce pulse widths that are no shorter than 2 ns. Given the emission

characteristics of UWB devices examined in this study, this may be inadequate for accurate 
6

In exceptional cases, the RF front-end is narrower than the IF section; those analyzers can

usually be retrofitted with wider bandwidth (essentially wide-open) front-ends. The result is that

the spectrum analyzer bandwidth is still ultimately limited by the IF section.

6-3

measurements of UWB pulse widths. But it will probably be more than adequate for

measurements of pulse repetition rate, pulse sequences, and gating behavior. It will also be

adequate for measurements of pulse-to-pulse dither intervals.

It is possible to utilize a wideband time domain measurement system to digitize a pulse

(preserving the phase information) and then transform the waveform into a wideband spectrum,

using the Fourier transform, as described in Section 5. Such a spectrum may in turn be convolved

with narrower IF bandwidths to replicate the spectrum envelope that would be measured in any

arbitrarily chosen receiver IF. While this approach may be feasible for some UWB systems, pulse

to pulse waveform variation may require repetitive sampling measurements to form a spectrum,

and dithering may make repetitive sampling difficult or impossible. Dynamic range may also be

limited for some pulse measurements.

6.2.2 Bandwidth Limited Frequency Domain Measurement Theory

The convolution of a frequency domain measurement may be limited by either the RF front-end or

the IF bandwidth of the measurement device, assumed to be a spectrum analyzer. The IF

bandwidth is normally the limiting factor.6 For this discussion, the spectrum analyzer convolution

function is assumed to be essentially the IF filter shape.

If the IF bandwidth curve is substantially narrower than the spectrum being measured, then the

convolution of the two functions is nearly identical to the input spectrum function. In this case,

the spectrum measurement is nearly identical to the spectrum function that was applied to the

analyzer input.

The features visible in the measured spectrum will only be resolvable down to the IF filter width.

Features narrower than the filter will convolve to yield merely the IF filter shape. To further

resolve those features, narrower IF filtering must be used.

While narrower IF filtering provides better resolution on spectrum features, two drawbacks 

result. The first is true for all spectrum measurements, while the second will occur in particular

cases. The first problem is that the measurement takes proportionally longer to complete as the IF

is narrowed. A trade-off results between measurement time and amount of detail in the resulting

spectrum measurement. Measurement efforts must balance these factors. In practice, the width of

the IF filtering will often be determined by the need to resolve the spectrum with the same

bandwidth as a particular type of potential victim receiver. For example, a 30 kHz bandwidth 
7The sinc2

 spectrum is only dominant through the first few lobes; more extended portions

of the spectrum are dominated by transient effects in the rising and falling edges of the pulses.

6-4

may be selected in the spectrum analyzer to match as nearly as possible the 25 kHz bandwidth of

some types of land mobile radio receivers.

The second problem is that the signal-to-noise ratio (SNR) of the measured signal can decrease in

narrower IF bandwidths. This will occur if the noise power in the measurement system decreases

more slowly in the IF bandwidth than the power coupled from the spectrum being measured. The

result is decreased dynamic range in the measurement. For example, in a narrow IF bandwidth the

spectrum might only be measured 20 dB down from the highest point, as compared to perhaps 35

dB down in a wider bandwidth.

To determine when dynamic range will decrease with narrow IF bandwidth, consider first the

manner in which measurement system inherent thermal noise varies as a function of IF bandwidth.

The noise power present is directly proportional to the IF bandwidth. Measured in decibels, the

inherent noise power therefore varies as 10 log (IF bandwidth).

Likewise, if the measured spectrum is noise, or approximates noise characteristics, then the

spectrum power convolved in the IF varies as 10log10 (IF bandwidth). In this case, the SNR of the

spectrum is constant with IF bandwidth, and the dynamic range of the measurement is unaffected

by the choice of IF bandwidth. A drawback to a narrower IF bandwidth is the longer time

required to complete the measurement.

If the measured spectrum is not noise-like, the convolved measured power may change at a rate

that is faster than 10log10 (IF bandwidth). This case occurs for spectra generated by pulsed

transmitters. Consider the case in which pulse width is t, pulse repetition interval is T, and

normalized voltage measured in a spectrum analyzer peak detector at the fundamental frequency

is A. For such transmitters, the spectrum that contains most of the transmitted power consists of

lines spaced (1/T) apart. The line power envelope is classically sinc2, with the first two nulls

occurring at ±(1/t) relative to the fundamental frequency and subsequent nulls occurring at

intervals of (1/t).7

The power measured in a line at the fundamental frequency is

(6.1)

and the line power in decibels is

(6.2)
6-5

The average power is

(6.3)

or, in decibels

(6.4)

where duty cycle = (t/T) .

For pulsed emissions, peak power is the rate at which energy is transmitted during each pulse.

This is therefore a linear function of the ratio of the pulse width to the pulse repetition interval.

Therefore, the peak power is related to the average power and the line power by the following

relationship:

(6.5)

For n lines convolved within a measurement bandwidth, the measured peak power varies with n as 

(6.6)

Since the number of lines within the convolution bandwidth is proportional to the bandwidth, the

peak power varies as 20log10 of the measurement bandwidth for line spectra. Figures 6.1 and 6.2

illustrate the behavior of a line spectrum convolved with bandwidths that range from significantly

less than the line spacing to significantly wider than the pulse width. For typical analogous UWB

device emissions, note that the width of the central lobe corresponding to that in Figure 6.2 will

be on the order of a gigahertz or more.
6-6

Figure 6.1. Detailed line spectrum measurement for a fixed-pulse-repetition rate

transmitter. Pulse width is 1 μs and pulse repetition rate is 10 kHz (pulse

repetition interval is 100 μs). Duty cycle is 10log(1/100) = -20 dB. Peak

power from the transmitter is -20 dBm and RMS average power is -40 dBm.

Measured line power is -60 dBm. Multiplying the power per line by the

number of lines in the central lobe of the spectrum at the 3-dB points gives [(-

60 dBm + 20 dB) = -40 dBm] for the computed average power, in agreement

with the known average power. Note 20log(bandwidth) progression for

measured power when convolution bandwidth exceeds the line spacing.
6-7

-80

-70

-60

-50

-40

-30

-20

-10

0

2490 2495 2500 2505 2510

prr = 10 kHz

pulse width = 1 μs

<= 3 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

2 MHz

Frequency (MHz)

Measured power relative to maximum convolved power (dB)

Figure 6.2. The spectrum of Fig. 6.1 is shown for wider measurement bandwidths.

20log(bandwidth) progression holds up to bandwidth of (1/pulse width)= 

1 MHz. When bandwidth exceeds (1/pulse width), only a small additional

percentage of power is convolved.

As long as measurement bandwidths are narrower than the convolved spectrum features, the

measured emission envelopes of all peak-detected emission spectra will vary as a function of

measurement bandwidth at a rate between 10log10(bandwidth), as for noise, and 20log10

(bandwidth), as for pulsed signals. Other signal modulations may produce rates intermediate

between 10log10 and 20log10. UWB device emission levels may vary with bandwidth at such

intermediate rates.

Measurements of UWB emissions will show the functional bandwidth dependence empirically. If

the dependence is 10log10 (the same as thermal noise in the measurement system), then the SNR

and dynamic range of the measurement will be constant as a function of measurement 

bandwidth. If the UWB emission dependence is between 10log10 and 20log10 of measurement

bandwidth, then SNR and dynamic range of a UWB spectrum will increase with increasing

measurement bandwidth, until the measurement bandwidth equals the emission bandwidth of the

UWB emitter. If measurement bandwidth exceeds the UWB emission bandwidth, the SNR and 
8As the emission bandwidths of UWB devices are often in excess of a gigahertz, it may be

assumed that a bandwidth limited measurement system will not have an IF bandwidth wide

enough to completely convolve the UWB emission.

9The results of measurements performed with these procedures are reproduced in detail in

Appendices A-E, and are summarized in Section 8 of this report.

10A vector signal analyzer (VSA) may be substituted for the spectrum analyzer and

oscilloscope specified in this list. However, ITS has not yet performed UWB emissions with a

VSA, and the ultimate feasibility of these measurements with a VSA has therefore not been

determined by this laboratory.

11For example, this capability may be provided by an Agilent (formerly Hewlett-Packard)

85685A preselector. Other COTS RF front-ends are available from other manufacturers; the

equipment described here is specified only for the purpose of explaining the type of capability that

the RF front-end is expected to contain.

6-8

dynamic range will decrease, as the convolved power in the UWB spectrum will remain constant

but the thermal noise convolved by the measurement system will increase at a 10log10 rate.8

6.3 Measurement Approaches

The application of hardware and software to the task of characterizing UWB emissions is

described below. Unless otherwise noted, the descriptions are taken directly from operational

measurements performed on UWB devices at the ITS laboratory.9

6.3.1 COTS Hardware Requirements

The following types of equipment are generally required for UWB measurements:10

• spectrum analyzer with general purpose interface bus (GPIB) capability,

• digital oscilloscope with GPIB capability,

• RF front-end hardware such as low noise amplifiers, bandpass filters, and

attenuators, packaged either as discrete components or within COTS RF frontend preselectors,11

• broadband antennas designed for electromagnetic interference (EMI) and

electromagnetic compatibility (EMC) measurements, and
12An example of this type of interface is the National Instruments GPIB-to-parallel

interface adapter. Other types of GPIB adapters are available from other manufacturers; this

device is specified only for the purpose of describing the type of capability required.

13Some spectrum analyzers with integrated APD capabilities are becoming available as of

the release date of this report. These machines were not available for use by ITS when the report

was written.

14An adequate measurement is defined to be efficient (requiring a minimum of operator

time and knowledge to perform), repeatable, and automatically documented by the measurement

system.

6-9

• laptop PC-compatible computer or equivalent that can be connected to a GPIB

or equivalent interface for instrument control and data acquisition.12

6.3.2 Software Requirements

Some measurements that characterize UWB emissions, such as amplitude-probability distributions

(APDs, described below), require the acquisition of thousands or millions of data points.13 Such

measurements are difficult or impossible to perform adequately14 with systems that are manually

controlled and operated. Raw measurement data may be so voluminous as to make analog

recording techniques (e.g., pen-plotter outputs) impractical; some types of data analysis (e.g.,

APD generation from raw data) operate on data quantities that can only be reduced by computers.

For these reasons, measurement laboratories engaged in UWB emission characterization should

be equipped with a computer that can be used to control measurement instrumentation, acquire

data through a widely available bus interface (e.g., GPIB), and analyze the resulting data.

Software to perform limited instrument control, data retrieval, and analysis will be necessary to

complete the measurements and produce analyzed data sets. The software should perform the

following functions:

• Set spectrum analyzer parameters, including center frequency and frequency

span, IF bandwidth, video bandwidth, detector type, sweep time, attenuation,

reference level, and number of points per data trace (if necessary);

• Set digital oscilloscope measurement parameters, including time base, voltage

scale, coupling mode, and trigger mode;

• Command a specified number of data traces (e.g., 1000 traces) to be triggered on

a spectrum analyzer; and
15At the ITS, the software used for these measurements was originally written to control a

custom-designed NTIA measurement facility, the Radio Spectrum Measurement System (RSMS).

As such, the RSMS software package is much more complex than is required for UWB device

emission measurements.

6-10

• Command a specified number of data traces to be taken on a digital oscilloscope;

• Record to selected media (hard disk, magneto-optical disk, etc.) all triggered data

traces from a spectrum analyzer and a digital oscilloscope to the controlling

computer. Each trace from the measurement instrument should be recorded as an

individual data record; and

• Record critical supporting information with each data record. Such

documentation might consist of instrument settings, date, time, and logistics

information such as the name of the device being measured and the measurement

engineer.

The software must support the following functions on recorded data:

• Add calibration factors to measured data values, if such corrections were not

included in raw data (necessary corrections are described in detail below);

• Correct measured spectra for variable effective aperture of the measurement

antenna, as described in Part 4 of Appendix C and (also below in this section);

• Retrieve individual data records (with corrections described above) for CRT

screen display and printing (these should include spectrum measurements and

time waveforms);

• Process hundreds or thousands of individual spectrum analyzer or oscilloscope

data records into APDs, as described below (software should store generated

APDs for later retrieval and printing). (Alternatively, some newly available

spectrum analyzers incorporate integrated APD capabilities. See footnote 13.)

Commercially available measurement and data analysis software packages that perform every

function described above probably are not available; however, some combination of commercially

available packages may be adequate to perform the necessary functions.15
6-11

1 meter

Anechoic enclosure

GPIB (data and control)

GPIB (data and control)

321.4 MHZ IF

RF

video

out

Optional

components

for wideband

(10 MHZ or

wider IF)

measurements

Spectrum

Analyzer

Digital

Oscilloscope

PC

Compatible

Computer

RF front end filter

(bandpass, YIG,

varactor, etc.)

Low noise

amplifier

Broadband

measurement

antenna

A

UWB

device

IF

Spectrum

Analyzer

Wideband IF

Log Amplifier

RF

A

Figure 6.3. UWB emission measurement block diagram. 

6.3.3 Generalized Measurement System

A generalized measurement system for characterization of UWB device emissions is shown in

Figure 6.3. The components shown in the figure are described in detail below.

Measurement Environment. The UWB device should be placed in an environment where

measurement system multipath and external radio signals are eliminated or minimized. The best
16Some commercially available antennas may perform adequately at frequencies beyond

their nominal specifications. Such performance extensions can be determined at a qualified

antenna test facility.

17The antenna must be located in the far field of the device being measured.

6-12

possible choice is a high-performance anechoic chamber. For ITS measurements, no such facility

could be scheduled for use within the time constraints of this project. Two alternative

environments that were utilized during ITS measurements were a large room (measuring about 15

m by 15 m by 10 m high) with a metal ground plane, and a smaller laboratory room with a

temporary enclosure built of stacked anechoic foam blocks. In the large room, UWB devices were

mounted on Styrofoam blocks about 2.5 m above the ground plane; in the anechoic enclosure,

UWB devices were mounted at a height of about 1 m above the floor.

Background signals may present a problem for UWB measurements not performed in an anechoic

chamber. This problem is exacerbated by the emission bandwidths of UWB signals, which may in

some cases occur in portions of the spectrum occupied by prominent background signals including

UHF television (512-806 MHz); analog cellular and trunked mobile communications (806-

902 MHz); pagers (928-932 MHz); PCS cellular communications (1.9 GHz); air route

surveillance radars (1215-1400 MHz); and tactical air surveillance radars (2.7-3.7 GHz, and

especially 2.7-2.9 GHz for airport surveillance radars). Signals from these sources may be

attenuated somewhat by the walls of the laboratory, and may be further reduced within an

anechoic foam enclosure. But measurements of background levels should be performed prior to

UWB device emission measurements, and can subsequently be subtracted from the measured

UWB spectra to yield the UWB spectra alone.

Measurement Antenna. Measurement antennas must have a sufficiently wide frequency

response. This may be a non-trivial problem for UWB device emissions. UWB measurements by

ITS have typically spanned such frequency ranges as 600 MHz to 6 GHz, a full octave. Multiple

antennas may be required if the frequency range to be measured is not completely included by a

single antenna.16 Commercially available wideband horn antennas have performed well for all ITS

and some National Institute of Standards & Technology (NIST) measurements described in this

report. If the measurement antenna polarization is linear, the antenna orientation needs to be

matched to the polarization orientation of the device being measured. This is accomplished by

rotating the measurement antenna until the received power level is maximized.

Because UWB device emissions may be relatively low-amplitude, the measurement antenna will

usually be placed as close as practicable to the device. The ITS measurements performed in the

course of this project utilized an antenna at a distance of 1 m from the devices being measured.17

The measurement antenna must be calibrated for the distance at which measurements are

performed. Again, wideband horn antennas have proven to be workable in this respect, as they 
18Higher noise figure results in loss of sensitivity; gain that is too low will fail to overdrive

measurement system noise, while gain that is too high will reduce available dynamic range. But

since UWB emission measurements typically observe low-amplitude signals, dynamic range is not

likely to be an important problem for these measurements.

19Overload may be checked by inserting attenuation between the antenna and the

preamplifier and observing a corresponding drop in signal level at the spectrum analyzer output. If

the drop is less than the attenuator value, then feed-through directly into the preamplifier or the

spectrum analyzer may be occurring, or the preamplifier may be in an overload condition.

20Spectrum analyzer noise figure at a given frequency may be checked by terminating the

input, setting RF attenuation to zero, IF bandwidth to 1 MHz, video bandwidth to 1 kHz, and

detector to sample. The difference between the measured noise level and -114 dBm is the analyzer

noise figure at that frequency.

6-13

typically can be acquired with calibration curves made at 1 m. If the measurement antenna does

not have a constant effective aperture (i.e., if the antenna gain does not increase as 20log10

[measurement frequency]), the resulting spectra must be corrected to the constant effective

aperture case. Section C.4 of Appendix C describes the necessary spectrum corrections for nonconstant effective aperture measurement antennas.

RF Front-end. Because UWB device emissions are often too weak to overdrive the noise figure

of spectrum analyzers, a low-noise amplifier may be required at the output of the measurement

receiver antenna. All devices measured by ITS for this report required such front-end

preamplification. In an optimized measurement system, the sum of the preamplifier gain and

preamplifier noise figure should be nearly equal to the noise figure of the spectrum analyzer across

the frequency range to be measured.18

Although UWB device emissions may have low amplitude within the convolution bandwidth of

the measurement system, the total power convolved by the front-end preamplifier may be high

enough to cause overload of that component. The preamplifier may also experience overload due

to ambient signals if the measurements are not being performed in an anechoic chamber. Overload

conditions must be checked by measurement personnel.19 If overload is experienced, then

appropriate RF filtering will be required between the antenna and the preamplifier.

Spectrum Analyzer. A spectrum analyzer may be used to measure both the bandwidth limited

spectra and time domain information (when operated in zero hertz span). Spectrum analyzers

typically have noise figures of 25 dB or more at frequencies below 2.5 GHz, and noise figures

increase with frequency due to mixer noise, typically at a rate of about 5 dB per mixing stage.20

As noted above, the preamplifier noise figure and gain should be optimized to work with this

noise figure.
21The IF was used at 10 MHz and 20 MHz for the ITS measurements.

6-14

Detector and Oscilloscope. Time domain measurements at bandwidths wider than those available

in the spectrum analyzer are performed with a diode detector and an oscilloscope, as shown in

Figure 6.3. For these measurements, the RF line is disconnected from the spectrum analyzer and is

routed through the diode to the oscilloscope. Note that the bandwidth limited video output from a

spectrum analyzer can also be routed to the oscilloscope, omitting the detector.

Optional Measurement Hardware for Bandwidths Wider than Nominal Spectrum Analyzer

IF. For the ITS measurements, a commercially available wideband IF module was procured for

one of the spectrum analyzers. This module was specified with performance to 100 MHz

bandwidth, but the spectrum analyzer RF front-end was only 22 MHz wide, thus limiting the

overall performance to a maximum of 22 MHz.21 Because the wideband IF section did not include

a detector, ITS measurements utilized a wideband log amplifier/detector at that module’s output.

The log detector output was connected to the oscilloscope input. Calibration was performed

manually for this project, although software will be required in the long term.

6.3.4 Calibration

Measurement devices must be individually calibrated by certified facilities. But additional

calibrations and corresponding corrections may be expected to be necessary for the complete

UWB measurement system. These include:

Corrections for Non-constant Effective Aperture Measurement Antenna, and Conversion

to Incident Field Strength. As described in Section C.4 of Appendix C, the measured spectra

must be corrected for non-constant aperture of the measurement antenna. This correction is also

required if the measured spectra are to be easily converted to incident field strength. Conversions

between measured power in a circuit and incident field strength are derived in Section C.1 - C.3

of Appendix C.

RF Front-end. The frequency-dependent gain of the RF front-end (including the loss through the

front-end filtering, if any, and the RF line from the preamplifier to the spectrum analyzer) must be

subtracted from the power values measured at the spectrum analyzer. ITS generally uses a Yfactor noise diode calibration procedure for this purpose. Signal generators may also be used.

Special Wideband IF Unit and Discrete Wideband Log Amplifier/Detector, if Used. The

wideband IF unit used by ITS for some measurements required provision for an IF gain factor

within that unit. The response of the discrete log amplifier/detector between the wideband IF and

the oscilloscope must also be calibrated, if these devices are used in the measurement system

(Figure 6.3).
22This is the case, for example, for some spectrum analyzers which couple input signals at

frequencies below two or three gigahertz directly into a mixer downconverter.

6-15

6.4 Types of UWB Measurements

6.4.1 Spectrum Measurements

Spectrum analyzer RF front-ends are sometimes on the order of two to three gigahertz wide for

tuned RF frequencies below about three gigahertz.22 At higher frequencies, such analyzers may

incorporate a tunable RF bandpass filter which may be approximately 20 MHz wide. Some

spectrum analyzers may be used in combination with external preselectors. These preselectors

typically utilize varactor filters at frequencies below 500 MHz and yttrium-iron-garnet (YIG)

filters at higher frequencies. The bandwidths of these preselector stages may be expected to be

between 15 MHz and 35 MHz. If the ultimate bandwidth limit is in the IF section, then the widest

bandwidths available will most often be between 3 MHz and 10 MHz. In exceptional cases,

spectrum analyzer IF sections may be procured that are as wide as 100 MHz (as in Figure 6.3).

Spectrum Envelope Measurements. ITS spectrum analyzer measurements of UWB emissions

were found to be most efficiently performed with the following spectrum analyzer settings:

Swept-frequency mode, zero front-end RF attenuation, reference level chosen as appropriate for

the UWB emission levels, positive peak detection, maximum-hold trace mode, video bandwidth at

maximum, and IF bandwidth selected progressively from the widest available bandwidth (3 MHz)

to a much narrower setting (10 kHz, for approximation of land mobile radio bandwidths).

Swept-frequency measurements have been found to be fast and practical for UWB device

emissions in this project. Only at bandwidths narrower than about 30 kHz does the sweep time

exceed a few seconds across the requisite several gigahertz of spectrum. For some spectrum

analyzers, it may be impossible to perform a single sweep across the entire range of interest (e.g.,

800 MHz to 6 GHz). In those cases, the spectrum is swept in two or more measurements (e.g.,

800 MHz to 2.5 GHz, and then 2.5 GHz to 6 GHz). Automation of these measurements by ITS

has increased the speed and reliability of the UWB measurement project and its results.

For spectrum measurements in IF bandwidths greater than the widest available in a conventional

spectrum analyzer (e.g., wider than 3 MHz or 10 MHz, depending upon spectrum analyzer

model), an optional wideband IF module and log amplifier/detector (or other circuitry with

equivalent operational capability) will be required, as shown in Figure 6.3. The wide-bandwidth

measurement is performed as follows: The spectrum analyzer is tuned to the first frequency to be

measured in the spectrum, in a zero hertz span. A single oscilloscope trace is acquired through 

the wideband IF and the log amplifier, as shown in Figure 6.3. The oscilloscope sweep time must

be long enough to ensure that a maximum-power pulse from the UWB device is recorded 
6-16

somewhere on the trace. After the oscilloscope trace is recorded, the spectrum analyzer is tuned

(still in a zero hertz span) to a second frequency. The second frequency should be as close as

practicable to the first frequency plus the IF bandwidth. For example, if the first frequency is

800 MHz and the IF bandwidth is 20 MHz, then the second frequency is 820 MHz. The

measurement is repeated, with a data trace recorded as before. This process continues until the

entire spectrum has been measured. For the example given here, if the spectrum to be measured is

800 MHz to 6 GHz, then the set of frequency steps at which measurements will be recorded from

the oscilloscope will be 800 MHz, 820 MHz, 840 MHz, etc., to 6 GHz. For the hypothetical case

described here, the measurement would require the acquisition of 260 oscilloscope traces; 

therefore, this process is most efficiently and reliably performed with an automated (softwarecontrolled) data collection system.

To generate the spectrum in the wide IF bandwidth, each oscilloscope trace must be retrieved

from storage, the maximum (peak-voltage) point on each trace must be determined, and those

voltages must be converted to power values in the measurement circuit. This is done by making

use of the log amplifier/detector calibration that was performed earlier. The final spectrum for the

example considered here would consist of 260 data points. The process of retrieving thousands of

raw data points, selecting the 260 maximum amplitude points from each trace, converting those

points to power in the measurement circuit, and plotting the points as a spectrum requires

software.

Bandwidth-dependence Measurements. As noted in 6.2.2 (above), measured emission

amplitudes from UWB devices vary as a function of measurement bandwidth. This variation

represents the relative levels that are coupled into receivers that are bandwidth limited. For

purposes of EMC/EMI characterization, and possibly also for regulatory and statutory purposes,

it is therefore necessary to understand the rate of the progression for extrapolation to arbitrarily

specified receiver bandwidths. Determination of this bandwidth dependence may be performed in

two ways. The first is to measure the entire spectrum (as above) in at least several bandwidths,

and note the progression between one spectrum and the next. The second method, which can be

performed at specific frequencies of particular interest (e.g., a global positioning system satellite

frequency at 1575 MHz) may be performed as follows:

Place the spectrum analyzer in a zero hertz span mode at the frequency of interest. Set the

detector to positive peak and the attenuation to zero. Set the video and IF bandwidths to their

widest possible values. Set the sweep mode to “single” (i.e., so that only one sweep will be

acquired when the proper trigger button on the analyzer is pushed). Set the sweep time to 60

seconds. With the UWB device emissions in progress, start the single sweep. At intervals of 6

seconds, reduce the IF bandwidth in a 1, 3, 10 progression or some approximate equivalent. The

result will be a stair-step trace as shown in Figure 6.4. This progression is a direct indication of

the relative peak-detected coupling at IF bandwidths of 3 MHz, 1 MHz, 300 kHz, etc., down to

100 Hz.
6-17

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F2R7.ep

Dimensionless Units

Received Power Relative to 3 M

Hz IF Bandwidth (dB)

Figure 6.4. Bandwidth progression measurement example for a UWB signal. The

emission is a pulse sequence (2 MHz pulse repetition rate) that dithers about

1% of the nominal pulse repetition interval on a relative (pulse-to-pulse) time

base. The variable-rate progression demonstrates the usefulness of empirical

data for determination of coupled UWB peak power as a function of receiver

bandwidth.

Depending upon the transmitter characteristics, it has been observed that the bandwidth

progression is not necessarily constant from one step to the next, as seen in Figure 6.4. Such 

non-uniformity is documented by this measurement technique. At wider bandwidths, the

progression indicates extrapolated levels for bandwidths wider than those that can be directly

measured (e.g., 50 MHz). The method of extrapolation is described in Section 8 of this report.
23Actually, percentage of samples, but if the sampling is not correlated with the device

emissions, this becomes equivalent to percentage of time.

6-18

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.001 .1 1 5 10 20 35 50 70 80 90 95 99

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slope

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o

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B

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d

width (d

B

m)

Figure 6.5. APD measurement example. The data for bandwidths of 10 kHz to 3 MHz

were acquired with a spectrum analyzer. The data in 10 MHz and 20 MHz

bandwidths were acquired with a commercially available wideband IF

spectrum analyzer, detected with an external log amplifier, and recorded from

a digital oscilloscope. Refer to Figure 6.3 for measurement system block

diagram.

6.4.2 Amplitude Probability Distribution Measurements

Amplitude probability distribution (APD) measurements show the percentage of time23 that

emissions from a device exceed a given power threshold. APD plots are typically produced on

Rayleigh scales, as shown in Figure 6.5. APDs have been a critically important tool for the ITS

characterization of UWB emissions. An APD curve will show the entire time-occupancy

distribution at a frequency and in a selected bandwidth. It may be processed to show peak level in

the bandwidth and several averages, including root-mean-square (RMS) (that is, linear average

power), average voltage, and log average. APDs are discussed in more detail in Appendix A.
24APD sampling may also be performed directly with some newly available spectrum

analyzers. Vector signal analyzers may also be feasible, but ITS has not tried a VSA to date.

25In the course of this project, it was observed that the sample detector bandwidths of

some spectrum analyzer models are narrower than the widest IF bandwidths in the same

analyzers. This problem can be checked as follows: Terminate the spectrum analyzer input, set the

frequency span to zero Hz, and set the tuned frequency to any value (e.g., 1 GHz). Select the

widest IF bandwidth with positive peak detection and maximum-hold trace mode. Allow the trace

to build to a maximum. Retain the resulting trace on the analyzer display. Select a second trace,

and change to sample detector mode. Select maximum-hold trace mode and allow the resulting

trace to build to a maximum. If the maximum-hold sample detector trace does not eventually build

to the same level as the maximum-hold positive peak detector trace, contact the spectrum

analyzer manufacturer.

6-19

APD measurements require that power level in the UWB device emission be sampled as a

function of time at the desired frequency and bandwidth. The time sampling may be performed

with either a spectrum analyzer or a digital oscilloscope.24 For either approach, the APD 

measurement will require automation of at least some portions of the measurement system.

APD Sampling with a Spectrum Analyzer. Unless a spectrum analyzer incorporates an

integrated APD capability (see footnote 13), APD sampling is performed as follows: The

spectrum analyzer is tuned in a zero hertz span to the desired frequency. Attenuation is set to

zero, and reference level is set to a value that is about 10 dB above the maximum UWB emission

amplitude. The spectrum analyzer video bandwidth is set to the widest possible value. Sample

detection is selected.25 The IF bandwidth is selected as required for the APD curve (in these

measurements, values between 10 kHz to 20 MHz were used). Sweep time is unimportant, with

the caveat that it must not be synchronized with the UWB device operation in any way.

With the UWB device in operation, a sweep is taken, and the data trace that results is stored

electronically for future retrieval. This process is repeated until an adequate number of samples

have been collected. If 1,000 points are generated within each spectrum analyzer sweep, then 100

sweeps will collect 100,000 points. It has been determined that between 100,000 and 1,000,000

points are required for adequate APD data on a variety of UWB transmitters. In the experience of

ITS on this project, the measurement system needs to be automated for the generation and

acquisition of the corresponding number of sweeps (between 100 to 1,000) if the measurements

are to be performed efficiently and reliably.

Because APD measurements do not reconstruct waveforms, APD sampling rates are not

constrained by the classical Nyquist rate of (2 x sampling bandwidth). Slower sampling is

permitted under the condition that the samples are statistically uncorrelated with UWB device 
26The Rayleigh plot shows decibel power on the ordinate vs. 10log10 [-ln(% samples/100)]

for samples exceeding the ordinate power.

27UWB spectrum measurements performed in this project did not generally result in clearly

defined 3-dB bandwidths. Sometimes 10 dB was difficult to define; 20-dB points were generally

definable.

6-20

operations (e.g., the UWB pulse gating interval should not be synchronized with measurement

equipment sampling).

APD Sampling with an Oscilloscope. An oscilloscope may be required for the collection of data

samples at IF bandwidths exceeding the maximum available in conventional spectrum analyzer

IFs. The ITS system, for example, utilized a specially procured (but commercially available)

wideband IF unit that is housed in a spectrum analyzer chassis but that does not incorporate the

spectrum analyzer detector functions (Figure 6.3). An external, discrete wideband log

amplifier/detector was used to send 10 MHz and 20 MHz IF data to a digital oscilloscope. The

oscilloscope was used to collect APD data in the same manner as the spectrum analyzer. The

oscilloscope must not be set in an envelope-detection or peak-detection mode or some other

mode that biases the samples toward higher values. Oscilloscope data traces are recorded

electronically for later retrieval and analysis. As for the spectrum analyzer, this is best

accomplished with software-automated data collection routines.

APD Data Analysis. APD raw data must be converted into finished APD curves. Unless a

spectrum analyzer contains an integrated APD capability (see footnote 13), there is no practicable

way to do this without software. APDs may be generated by separating the points into

predetermined bin sizes (thus reducing the size of the finished file), or by accumulating all of the

100,000 or 1,000,000 points into a statistically indexed array (resulting in a larger final APD file

size). The APD curve is conventionally graphed on a Rayleigh plot,26 as shown in Figure 6.5.

APD curves may be further analyzed to determine the following critical emission parameters at

the measured frequency in the measured bandwidth: peak power, RMS average, average voltage,

and log average. ITS engineers have found APD curves to be useful in part because of the amount

of information that can be directly determined from them.

6.4.3 Pulse Width and Shape Measurements

Pulse Width. Detailed pulse width measurements require the use of a measurement systems such

as those described in Section 5. But COTS equipment may be used to estimate pulse width. Two

methods are available. The first is to measure the width of the UWB spectrum at 10 dB or 20 dB

points.27 The inverse is approximately equal to pulse width. This is only an approximate measure 
6-21

of emitted pulse width, and includes the effects of final-stage UWB RF components such as

output bandpass filters and the radiating antenna.

The second pulse width measurement method is to send the output of the measurement system to

a discrete-component wideband diode detector, and connect the detector output to an

oscilloscope. The oscilloscope may be used to estimate pulse width down to at least the inverse of

the maximum oscilloscope bandwidth in single-shot (as opposed to repetitive sampling) mode.

For a 50 MHz bandwidth, for example, pulse width may be estimated down to 20 ns. A scope that

operates at 1 GHz bandwidth without repetitive sampling may be adequate to measure pulse

widths down to about 1 ns. If scope bandwidth in single-shot mode is inadequate for measurement

of a UWB pulse width, then the first method (estimating pulse width from spectrum width) may

be the only available alternative.

Pulse Shape as a Function of Receiver Bandwidth. For purposes of EMI/EMC evaluation with

various receiver types, the shape of the pulse should be measured through a spectrum analyzer in

a variety of IF bandwidths. The pulse shapes are recorded in a range of bandwidths (e.g., 10 kHz

to 20 MHz) for future reference in studies of compatibility with receivers having corresponding IF

bandwidths. (This assumes that the pulse shapes so measured are determined by the impulse

response of the receiver IF section.)

6.4.4 Pulse Repetition Rate, Dither Sequence, and Gating Measurements

In order to measure nominal repetition rates, dither sequences, and gating, pulse sequences must

be measured. These have been found to be most easily measurable with the RF from the UWB

transmitter connected to a wideband discrete-component diode detector. The diode output is

connected to an oscilloscope and the pulse sequences are measured and recorded at time intervals

ranging from several seconds to as little as (1/oscilloscope bandwidth).

Nominal Pulse Repetition Rate (PRR). The nominal PRR, if any is present, is easily read from

an oscilloscope display.

Dither Measurement. Two types of dithering have been observed in UWB devices measured by

ITS during this project. One is absolute time-base dither, in which pulses are triggered relative to

an absolute time base. The other is relative time-base dither, in which each pulse is triggered at an

interval that is measured relative to the preceding pulse, with no reference to a fixed time base.

Absolute dither systems may have 50% dither on a pulse-to-pulse basis. Relative time-base

systems may have only a 1% dither from pulse to pulse, but the variance in pulse occurrence

grows with time. Both types of dither act to suppress lines in resulting emission spectra. 

Dither may be observed and documented by triggering a storage oscilloscope on a pulse and then

observing the intervals at which subsequent pulses occur. If absolute time-base dithering is

occurring, only a few pulses need be measured to document this behavior. If relative time-base
6-22

dithering is occurring, then a storage oscilloscope is used to trigger on a pulse, and then the

display is scrolled for an appropriate length of time (possibly many tens of milliseconds) to

observe long-term growth in the dither variance. That is, if the relative time-base dither is 1%,

hundreds of pulses must be observed after the trigger pulse to observe the total growth in the

variance of pulse-occurrence time.

Gating. Pulse gating is a higher-level modulation of the UWB output, in which the transmitter is

quiescent for intervals that are long compared to the nominal pulse repetition interval. Gating is

easily observed on an oscilloscope output. It can also usually be measured on a spectrum analyzer

in a zero hertz span mode, since the gating intervals are usually on the order of a few milliseconds

to tens of milliseconds in duration.

6.4.5 Peak Power Measurements 

Peak power may be measured in a bandwidth as described in Section 6.4.3 above. A spectrum

analyzer is used to determine peak power in either a spectrum trace or at a single frequency in a

zero hertz span mode. Peak power may also be read from an APD curve, as described above.

A significant problem is to measure total peak power in the emission bandwidth of the UWB

device. This is not generally feasible in a direct measurement with COTS equipment. ITS

equipment is currently capable of measuring peak power in 20 MHz bandwidth at the widest.

Although an upgrade to 50 MHz and ultimately 100 MHz is planned for ITS systems, the COTS

systems that will accomplish this are known to be expensive, and will require additional

calibration routines (requiring dual implementation in software and hardware).

In short, for many laboratories engaged in UWB emission measurements, IF bandwidths between

3 MHz to 10 MHz may be the widest available for the immediate future. And in no case will peak

power likely be measured at bandwidths equaling UWB emission bandwidths. Extrapolation to

wider bandwidths will be a necessity if wider-bandwidth peak power measurements are required.

Extrapolation will require the measurement of the emission in a succession of bandwidths (as the

stair-step measurement in Figure 6.4), so that the rate of progression can be measured.

6.4.6 Average Power Measurements

Two averages that are widely used for various purposes are root mean square (RMS) average and

logarithmic average. RMS average is the equivalent constant-rate of flow of energy from the

transmitter over a time interval that is long compared to the nominal pulse repetition interval.

Logarithmic average (average decibels) is required by such regulatory standards as Part 15 of

CFR 47.209. While the logarithmic average is sometimes used vernacularly in an 

interchangeable sense with the term “average power,” it must be stressed that the average decibel 
28The measurement of RMS power from UWB devices with power meters was not part of

this project.

6-23

level emitted by a transmitter is not equivalent to the RMS (arithmetic mean) average output

power level.

Measurement of RMS Average. At least five alternative techniques for using COTS equipment

to measure RMS emissions from UWB transmitters have been found to be feasible, as detailed

below.

RMS Detector Integrated into a Spectrum Analyzer. As of 2000, most

commercially available spectrum analyzers do not incorporate RMS detectors. But

as of late 2000 and early 2001, some analyzers with integrated RMS detectors are

becoming available on the commercial market. One such analyzer became available

to ITS engineers for evaluation purposes just prior to this report’s release, too late

to be used in the UWB measurements contained in this report. But ITS engineers

were able to evaluate the detector’s performance characteristics on some UWB

signals. The outcome of that preliminary evaluation indicated that this integrated

detector appeared to offer a technically feasible solution to the problem of

measuring RMS values of UWB emissions. It is anticipated that similar RMS

detectors in other newly available spectrum analyzers will be similarly feasible for

measurement of UWB average emissions. 

Power Meter RMS Measurement.28 RMS power may be measured directly with

a power meter, under some conditions. The power meter must utilize a bolometertype sensing head. The UWB signal must be at least 10 dB above the thermal noise

in the measurement head, so that the power being averaged is primarily an

indication of the UWB device emission and not of the thermal noise in the power

meter. The limited experience that ITS has in this particular measurement on

UWB-like signals has indicated that it is difficult to realize a sufficiently high SNR

on UWB signals if measurements are performed on radiated (as opposed to

hardline-coupled) signals. Diode-type power meters may have insufficient dynamic

range for the required UWB measurements.

Spectral Line Power Measurement. If the UWB device produces a line spectrum

(i.e., the signal is not a dithered pulse sequence or other noise emission), then

average power may be derived from measured power in a maximum-amplitude

spectrum line. The functional relationship is given by Equation 6.5, above.
29This includes devices that generate line spectra; for example, absolute time-based dither

signals generate lines at the interval of a PN code sequence, on the order of 1 kHz or 10 kHz.

6-24

APD Curve Calculation. RMS may be directly computed from the APD curve for

the frequency and bandwidth in question. It may also be computed as the APD is

generated.

Computation from Spectral Line Measurement or from Duty Cycle and Peak

Power Measurement. The peak power is related to the average power and the

power in a maximum-amplitude line by the relationship in Equation 6.5. However,

many UWB devices do not generate line spectra.29 For these devices, average

RMS power may be calculated by multiplying the peak power in a bandwidth by

the duty cycle. Duty cycle may be measured directly (if pulse width can be

measured directly or at least estimated (see above)). Less desirably, from the

standpoint of measuring quantities directly, duty cycle may be taken from a

specification sheet for the device.

Measurement of Logarithmic Average. A logarithmic average may be measured directly from a

spectrum analyzer as follows: With the analyzer in a zero-hertz span mode, set IF bandwidth to a

desired value (e.g., 1 MHz, if log average power in 1 MHz is required). Video bandwidth is

narrowed to a value on the order of a few hertz. (10 Hz to 10 kHz video bandwidth is specified

by some regulatory measurement procedures.) The resulting log (decibel) average is read from the

spectrum analyzer display, with some caveats.

The caveats are critical, from a technical standpoint. First, ITS has discovered that some UWB

emissions generate log averages that are below measurement system noise, as shown in Figure 6.6.

In such a case, the attempted log average measurement converges on the measurement system

noise floor rather than the log average emission of the UWB device. Given that the measurement

system noise in the widely used COTS preselector was eleven decibels above ambient thermal

noise in this case, and that the broadband horn measurement antenna was placed 1 m from the

UWB device, it is difficult to design a COTS measurement system that will achieve better SNR on

the UWB signal than that shown in Figure 6.6. The failure of the log average technique to indicate

only the measurement system noise in this case means that the technique is not generally

applicable to UWB devices as a class. 

Log average may also be computed from an APD curve, just as for RMS average. But if the log

average is below the measurement system noise, the log average will not be computable from the

APD curve. 
6-25

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Figure 6.6. Log average measurement for a gated UWB transmitter with a low duty cycle. IF

bandwidth = 3 MHz and video bandwidth is narrowed from 1 MHz to 10 Hz ((a) through (f)).

This measurement converges on the measurement system noise floor, and thus fails to indicate the

device’s true log average. Measurement antenna was 1 m from transmitter, and measurement

system noise figure was 11 dB. Given that little improvement in signal to noise ratio would be

achievable for this measurement with COTS equipment, this serves as an example of the difficulty

in assessing average power from all types of UWB transmitters using this technique.
6-26

6.5 Summary of Bandwidth Limited Measurement Techniques for UWB Devices

In summary, a UWB measurement system utilizing COTS equipment for bandwidth limited

measurements can be achieved with commonly available laboratory hardware. The general

measurement configuration may be as shown in Figure 6.3. Low-noise preamplification will often

be required. Calibration of measurement equipment is critical, and should include corrections for

non-constant aperture measurement antennas.

The primary measurement equipment required includes a spectrum analyzer, an oscilloscope, a

discrete-component diode detector, a laptop computer that can perform simple commands and

data acquisition on the analyzer and the oscilloscope, and software that can perform limited

commands, data retrieval, and data processing as described above.

Emission spectra, bandwidth dependence of emission amplitudes, amplitude probability

distributions, pulse repetition rates, pulse dither sequences, and emission gating may be measured.

Pulse shapes may be measured as a function of receiver IF bandwidth. Pulse width may be

estimated but probably not measured precisely. Peak power in a bandwidth may be measured, and

may be estimated for any bandwidth. Average power measurements are more problematic; RMS

may be measured directly with an appropriately equipped spectrum analyzer or a power meter (if

important conditions are met). Average power may alternatively be estimated with alternative

methods. Log average may be directly measured for some UWB devices, but cannot be measured

for UWB emitters as a general class.
1The author is with the Institute for Telecommunication Sciences, National

Telecommunications and Information Administration, U.S. Department of Commerce, Boulder,

CO 80305. 

7-1

SECTION 7: EFFECTS OF TWO UWB SIGNALS ON THREE FEDERAL RADARS

Brent Bedford1

7.1 Introduction

Some UWB signals appear to occupy a relatively large portion of the spectrum when compared to

the spectral occupancy of the signals of conventional systems. This leads to the notion that UWB

systems may need to share the same spectrum with conventional incumbents. As in the case

where two conventional systems share spectrum, questions are raised about the effects of a new

system sharing spectrum with an incumbent. NTIA conducted the following study to determine

the levels at which a UWB signal could be present within a radar receiver.

The study involved two types of UWB signals. Both types of UWB signals consisted of pulses at

a 10 MHz pulse repetition rate (PRR). Both types had the same pulse shape and amplitudes. The

first signal type was dithered using pulse position modulation over 50% of its basic repetition rate. 

In other words, there was a basic repetition interval of 100 ns. Each pulse was delayed from 0 ns

to 50 ns within each interval. The position for each pulse was determined by a uniformly

distributed random number generator. The second signal type was not dithered; the pulses were

generated at a constant rate. The non-dithered signal produces spectral lines which have a

frequency spacing equal to the reciprocal of the pulse rate. Since the pulse rate used in this study

was greater than the receiver's bandwidth, for the case of the non-dithered signal, only a single

spectral line was present within the receiver's passband.

Three Federal Aviation Administration (FAA) radio receivers were involved in the testing: the Air

Route Surveillance Radar (ARSR-4), the Airport Surveillance Radar (ASR-8) and the Air Traffic

Control Beacon Interrogator (ATCBI-5). The ARSR-4 is a long range radar that detects targets

up to 518 kilometers away. The ASR-8 assists with traffic control at airports and detects targets

up to 124 kilometers away. The ATCBI-5 transmitter interrogates transponders that are located

on aircraft, while its receiver detects and processes the responses from the aircraft transponders. 

All three of these receivers are located at the FAA Mike Monroney Aeronautical Center (MMAC)

in Oklahoma City, Oklahoma. 

7.2 Radiated Measurements

The radiated measurements involved radiating a signal over a line-of-sight path from the UWB

source to the receiver under test. The levels, relative to the receiver's noise floor, were measured
7-2

and recorded. The results of this study will assist with determining any effects on these receivers

if their spectrum is shared with UWB devices. The radars at the MMAC are good representations

of radars that are operating in the field. These radars are fully functional but they are not being

used to generate the critical information needed by air traffic controllers. The receivers are

connected to rotating antennas that are the same as the field units. The FAA uses these radars to

test new concepts before implementing the concepts on all their units and to train new personnel.

To conduct this study, the following approach was implemented. A measurement vehicle was

outfitted to record the received signal level from an operating radar while the vehicle was in

motion. A GPS unit was installed in the vehicle. The measurement started on the road next to

the radar. The starting location was stored as a way point in the GPS receiver. The GPS receiver

was set to display the distance and bearing from the vehicle to the radar. The vehicle was driven

along roads that approximated radials from the transmitter. When a peak in the signal level

occurred, the distance and bearing to the radar were recorded.

The results from the various runs were examined to identify the locations in which the maximum

signal level was received. By reciprocity, these locations would offer the best chances for seeing

any effects that the UWB transmitter would have on the receivers. The results showed that the

signal level right under the radar was not the maximum level. Radar antennas typically

concentrate their radiated energy into narrow beams. This is necessary for them to determine

bearing and range to a target. These beams are directed into the sky to detect airborne craft. The

lowest angle of interest for the beam is usually slightly above the horizon. The area under and

close to the antenna is not directly illuminated by the beam, so only a moderate signal level exists. 

An example of this is shown in Figure 7.1. For approximately the first 1.75 horizontal divisions,

the signal is relatively undefined with rapid variations. After 1.75 horizontal divisions, the spikes

are well defined and vary smoothly.
7-3

-20

-10

0

10

20

30

0.92 1.04 1.21

0.6

0.52

0.29

0.17

Received Signal Level (dB

m)

Figure 7.1. ARSR-4 signal level along a southern radial.

Foliage and buildings are other factors that influence the received signal level. When either of

these obstructions enter the line-of-sight path between the vehicle and the radar, the received

signal level drops rapidly and a good sample at that distance is lost. The measurements were

conducted over several approximately radial paths in an attempt to get samples over a large

number of distances.

Once the locations of highest signal level were known, the next part of the study took place. The

Intermediate Frequency (IF) output of the radar receiver under test was connected to the

spectrum analyzer. The noise level on the spectrum analyzer was noted at the expected IF

frequency before connection to the IF output. All the levels were measured with the spectrum

analyzer resolution bandwidth set to 1 MHz and the video bandwidth set to 10 Hz. This is the

same spectrum analyzer settings that are used in a FCC Part 15 compliance measurement for

frequencies above 1 GHz. The spectrum analyzer was connected to the receiver's IF output and

the noise level was noted again. It was necessary for the noise coming from the IF receiver to

raise the noise floor of the spectrum analyzer a perceptible amount in order for the measurements

to be valid. This was the case for all of the receivers in this study.

The measurement vehicle was configured to house the UWB transmitter. A ridged horn antenna

was mounted on a telescoping mast on the vehicle. While measurements were being conducted,

the horn antenna was elevated above the vehicle roof to prevent reflections. For each

measurement, the UWB transmitter was adjusted, by a variable attenuator, to produce an effective

isotropic radiated power of -41 dBm. This level corresponds to the radiated emission limits found

in FCC Part 15.209 for frequencies above 1 GHz.
7-4

The measurement vehicle was parked near some of the locations of highest signal level as well as

a few others. It usually was not possible to pull off the road right where a peak occurred. The

horn antenna was raised and pointed at the radar under test and the UWB transmitter was

activated. The radar transmitter was then turned off and its antenna was manually pointed in the

azimuth of the vehicle. The antenna was swung through many degrees to find the peak delta

marker value, which was recorded. The delta marker value consists of setting a reference level

with the UWB transmitter off, then measuring the decibel increase in the level with the UWB

transmitter on. Various attenuator settings were tried. The attenuator settings and the resulting

delta marker values were recorded.

In addition to the UWB transmitter measurements, a few incidental radiators were activated, with

line-of-sight paths to the radars, at the above sites. The delta marker values due to these devices

were recorded. This shows a comparison between UWB sources and incidental radiators.

7.2.1 ARSR-4 Radiated Measurements

The ARSR-4 receiver frequency was 1241.47 MHz. The PRR for the non-dithered UWB signal

had to be adjusted to put a spectral line within the passband of the receiver. It was calculated that

if the PRR was increased by 10.01 kHz, this would put a spectral line at 1241.241 MHz. This

new PRR was used for this radar.

The vehicle was driven along two paths. One path was along a southern radial that started at the

radar. The received signal level along this path is shown in Figure 7.1. The graph is labeled with

the distances from the radar in kilometers where the peaks occurred. The second path was on a

southern path that started 0.64 kilometers west of the radar. The received signal level along this

path is shown in Figure 7.2. On this graph, there is a gap in time, because the vehicle was

stopped and some time passed before the recording resumed. Although the vehicle was driven at

a constant speed and in most cases distances can be interpolated between the marks on the graph,

one should not interpolate across the gap in time.
7-5

-50

-40

-30

-20

-10

0

10

20

30

Gap in time

2.77

2.61

2.35

1.04

1.69

1.58

1.19

0.87

0.7

Received Signal Level (dB

m)

Figure 7.2 ARSR-4 signal level along a southern path.

Next, the vehicle was driven to three sites where measurements were performed. The first site

was 1.26 kilometers from the radar at a bearing of 178 degrees. For a 10 MHz PRR dithered

signal, radiated at the FCC Part 15.209 radiated emission limit, the delta marker value was 9 dB. 

Attenuating the UWB transmitter output by 6 dB brought the delta marker value down to 6 dB. 

Attenuating the UWB transmitter output by 12 dB brought the delta marker value down to 2 dB. 

Attenuating the UWB transmitter output by 15 dB brought the delta marker value down to 1 dB. 

For the non-dithered signal, radiated at the FCC Part 15.209 radiated emission limit, the delta

marker value was 0 dB. In this case, the UWB signal did not affect the noise floor. An electric

drill and an electric shaver were turned on to see if their radiation could be observed at the

receiver's IF. The electric shaver produced asynchronous spikes 10 to 15 dB above the noise

floor but this did not affect the noise floor. This is shown in Figure 7.3. The electric drill did not

affect the noise floor.
7-6

-85

-80

-75

-70

-65

-60

-55

0 0.002 0.004 0.006 0.008 0.010 0.012 0.014 0.016 0.018 0.020

Time (seconds)

Receiver IF (dB

m)

Figure 7.3. Asynchronous spikes produced by the electric shaver.

The second site was 2.08 kilometers from the radar at a bearing of 199 degrees. For a 10 MHz

PRR dithered signal, radiated at the FCC Part 15.209 radiated emission limit, the delta marker

value was 11 dB. Attenuating the UWB transmitter output by 6 dB brought the delta marker

value down to 8 dB. Attenuating the UWB transmitter output by 12 dB brought the delta marker

value down to 4 dB. Attenuating the UWB transmitter output by 18 dB brought the delta marker

value down to 2 dB. For the non-dithered signal, radiated at the FCC Part 15.209 radiated

emission limit, the delta marker value was 0 dB. In this case, the UWB signal did not affect the

noise floor. An electric drill and an electric shaver were turned on to see if their radiation could

be observed at the receiver's IF. Neither item affected the noise floor.

The third site was 3.17 kilometers from the radar at a bearing of 53 degrees. For a 10 MHz PRR

dithered signal, radiated at the FCC Part 15.209 radiated emission limit, the delta marker value

was 8 dB. Attenuating the UWB transmitter output by 6 dB brought the delta marker value down

to 5 dB. Attenuating the UWB transmitter output by 12 dB brought the delta marker value down

to 2 dB. Attenuating the UWB transmitter output by 15 dB brought the delta marker value down

to 1 dB. For the non-dithered signal, radiated at the FCC Part 15.209 radiated emission limit, the

delta marker value was 0 dB. In this case, the UWB signal did not affect the noise floor. An

electric drill and an electric shaver were turned on to see if their radiation could be observed at the

receiver's IF. Neither item affected the noise floor.

For a dithered UWB signal, radiated at the FCC Part 15.209 radiated emission limit, the 

maximum delta marker value was 11 dB. For a non-dithered UWB signal, radiated at the FCC
7-7

-50

-40

-30

-20

-10

0

10

20

0.17 0.25 0.3 0.4 0.52

0.1

Received Signal Level (dB

m)

Figure 7.4. ASR-8 signal level along a mostly northern path.

Part 15.209 radiated emission limit, the maximum delta marker value was 0 dB.

7.2.2 ASR-8 Radiated Measurements

To find the locations of highest signal level, the vehicle was driven along three paths. One path

started at the radar, went west for 0.22 kilometers, then went north. The received signal level

along this path is shown in Figure 7.4. The second path was on a northern radial that started

approximately 0.48 kilometers northeast of the radar. The received signal level along this path is

shown in Figure 7.5. The third path was a northward extension of the first path. This path

started 0.78 kilometers northwest of the radar and went north. The received signal level along

this path is shown in Figure 7.6. On this graph, there is a gap in time, because the vehicle was

stopped and some time passed before the recording resumed. Although the vehicle was driven at

a constant speed and in most cases, distances can be interpolated between the marks on the graph,

one should not interpolate across the gap in time.
7-8

-50

-40

-30

-20

-10

0

10

20

30

1.75

1.37

1.12

0.93

0.75

Received Signal Level (dB

m)

Figure 7.5. ASR-8 signal level along a northern radial.

-50

-40

-30

-20

-10

0

10

20

Gap in time

1.26

1.17

1.06

0.93

0.86 1.04

Received Signal Level (dB

m) Figure 7.6. ASR-8 signal level along a northern extension of figure 7.4.
7-9

Next, the vehicle was driven to two sites where measurements were performed. The two sites

were located as close to a recorded signal peak as possible. The first site was 0.4 kilometers from

the radar at a bearing of 326 degrees. For a 10 MHz PRR dithered signal, radiated at the FCC

Part 15.209 radiated emission limit, the delta marker value was 6 dB. Attenuating the UWB

transmitter output by 3 dB brought the delta marker value down to 3 dB. For the non-dithered

signal, radiated at the FCC Part 15.209 radiated emission limit, the delta marker value was 5 dB. 

Attenuating the UWB transmitter output by 3 dB brought the delta marker value down to 2 dB. 

An electric drill, electric hair dryer and an electric shaver were turned on to see if their radiation

could be observed at the receiver's IF. These items did not affect the noise floor.

The second site was 1.41 kilometers from the radar at a bearing of 11 degrees. For a 10 MHz

PRR dithered signal, radiated at the FCC Part 15.209 radiated emission limit, the delta marker

value was 1 dB. For the non-dithered signal, radiated at the FCC Part 15.209 radiated emission

limit, the delta marker value was 3 dB. Attenuating the UWB transmitter output by 2 dB brought

the delta marker value down to 2 dB. An electric drill, electric hair dryer and an electric shaver

were turned on to see if their radiation could be observed at the receiver's IF. These items did not

affect the noise floor.

For a dithered UWB signal, radiated at the FCC Part 15.209 radiated emission limit, the maximum

delta marker value was 6 dB. For a non-dithered UWB signal, radiated at the FCC Part 15.209

radiated emission limit, the maximum delta marker value was 5 dB.

7.2.3 ATCBI-5 Radiated Measurements

This radar at this location is mounted on top of a 30.5 meter tower. Because the typical ASR

beacon installation is not this high, these measurements may give rather optimistic results. If the

UWB signal was seen by this configuration, then the typical configuration would probably see the

signal at a higher level.

The vehicle was driven to two sites where measurements were performed. The first site was 1.26

kilometers from the radar at a bearing of 178 degrees. For a 10 MHz PRR dithered signal,

radiated at the FCC Part 15.209 radiated emission limit, the delta marker value was 0 dB. For the

non-dithered signal, radiated at the FCC Part 15.209 radiated emission limit, the delta marker

value was 0 dB. An electric drill and an electric shaver were turned on to see if their radiation

could be observed at the receiver's IF. These items did not affect the noise floor.

The second site was 0.5 kilometers from the radar at a bearing of 178 degrees. For a 10 MHz

PRR dithered signal, radiated at the FCC Part 15.209 radiated emission limit, the delta marker

value was 1 dB. For the non-dithered signal, radiated at the FCC Part 15.209 radiated emission

limit, the delta marker value was 0.6 dB. An electric drill and an electric shaver were turned on to

see if their radiation could be observed at the receiver's IF. These items did not affect the noise

floor.
7-10

For a dithered UWB signal, radiated at the FCC Part 15.209 radiated emission limit, the maximum

delta marker value was 1 dB. For a non-dithered UWB signal, radiated at the FCC Part 15.209

radiated emission limit, the maximum delta marker value was 0.6 dB.

While the vehicle was parked 15.2 meters from this radar, an electric drill and an electric shaver

were turned on to see if their radiation could be observed at the receiver's IF. These items did not

affect the noise floor.

7.3 Summary

It is instructive to gather the results into tables for examination. Table 7.1 contains the delta

marker values for the dithered UWB signal. Table 7.2 contains the delta marker values for the

dithered UWB signal. All of the values correspond to the cases where the UWB signal is radiated

at the FCC Part 15.209 radiated emission limit. The results indicate that all three radars

experience a noticeable effect for at least one and in some cases, both types of UWB signals that

were generated in this study. The ARSR-4 presented an interesting case in two ways. Within the

constraints of three site testing, Table 7.1 shows a relatively large effect with a dithered signal and

Table 7.2 shows no effect with a non-dithered signal.

Table 7.1. Delta Marker Values in Decibels for the Dithered UWB Signal.

Radar First Site Second Site Third Site

ARSR-4 9 11 8

ASR-8 6 1

ATCBI-5 0 1

Table 7.2. Delta Marker Values in Decibels for the Non-Dithered UWB Signal.

Radar First Site Second Site Third Site

ARSR-4 0 0 0

ASR-8 5 3

ATCBI-5 0 .6
1The author is with the Institute for Telecommunication Sciences, National

Telecommunications and Information Administration, U.S. Department of Commerce, Boulder,

Colorado 80305

8-1

8. MEASUREMENT SUMMARY AND CONCLUSIONS

Robert .J. Matheson1

8.1 Introduction

This section contains a summary of the measurements made on various UWB devices using the

measurement techniques described in Sections 5 and 6. The data presented in this section have

been selected from the total measurement set made on each UWB device; additional

measurement data for each device is included in Appendix D. 

Measurements from six devices are included.

Device A ................................... Section 8.3.1, Appendix D-A

Device B ................................... Section 8.3.2, Appendix D-B

Device C ................................... Section 8.3.3, Appendix D-C

Device D ................................... Section 8.3.4, Appendix D-D

Device E .................................... Section 8.3.5, Appendix D-E

Electric drill ................................. Section 8.3.6, Appendix D-F

8.2 Examples for Detailed Analysis

This section describes the selected measured data in detail, using examples from Devices A, B,

and D. It is expected that the reader will apply the explanations to corresponding data from the

remainder of the UWB devices, as appropriate. Data for each device will be presented in the

same order to facilitate locating data and comparing results between different devices

8.2.1. Device Description 

This is a short description of the UWB technology employed. ITS is making no claims that the

UWB device works as intended or as described. In most cases, ITS did not functionally test the

UWB device performance in any way (except to ascertain that the UWB transmitter was

apparently functioning), and consequently ITS is unaware of whether a specific UWB device

achieves any of its intended or claimed functional performance objectives. 
8-2

-2.5

0

2.5

0 5 10 15 20 25

Time (nanoseconds)

Volts

Figure 8.1. Device A full bandwidth pulse shape.

Device A uses a 10-kHz pulse repetition rate (PRR), apparently not gated, dithered, or modulated. 

ITS had no information on the operational aspects of this device (e.g., whether there was any

adaptive modification of transmitter modes depending on device location), and testing proceeded

on the assumption that the on-off switch was the only significant test variable.

8.2.2. Full-Bandwidth Pulse Shape

Figure 8.1 is an example of a full-bandwidth pulse shape measurement, as described in Section 5. 

Major objectives of this measurement include understanding full bandwidth pulse shapes,

evaluating the possible utility of various pulse-width models for engineering and regulatory

purposes (especially to relate pulse shape to the emission spectrum), and furnishing raw data for

fast Fourier transform (FFT) computations of emission spectra.

Depending on the UWB device, the full bandwidth pulse shape measurements were made on a

conducted basis and/or a radiated basis at a distance of 1 meter. Conducted measurements were

calibrated in dBm; radiated measurements were calibrated in voltage available at the terminals of

a ridged horn or TEM horn antenna at a distance of 1 m. For the radiated measurements, the

effects of the measurement antenna frequency-dependent gain and delay on the pulse shape have

not been corrected, though techniques to provide a calibration in absolute field-strength are still

being investigated. This process would involve performing an FFT of the uncorrected pulse,

correcting the resulting spectrum with frequency-dependent antenna delay and gain factors, and

performing a inverse FFT on the corrected spectrum to obtain a corrected pulse shape.
8-3

-80

-60

-40

-20

0

4000 5000 6000 7000

Frequency (MHz)

dBm

Figure 8.2. Device A conducted FFT spectrum, )f = 16.7MHz.

The Device A full bandwidth pulse shape in Figure 8.1 is relatively complex, involving multiple

lobes and many crossings of the zero-axis. A complex waveform like this is typical of UWB

signals that have been filtered with relatively high-Q resonant circuits to provide a signal limited

to a well-controlled RF bandpass. Other UWB devices (e.g., Device E) may have a much simpler

pulse shape with only a few zero-crossings.

8.2.3. FFT Emission Spectrum

Figure 8.2 is an example of an emission spectrum calculated via FFTs from the full-bandwidth

pulse shape in the previous figure. Device A’s full bandwidth pulse shape was measured in the

conducted mode, so the spectrum calculated via FFTs was calibrated in peak dBm in a specific

bandwidth = )f. In this case, a )f bandwidth of 16.7 MHz was used. For large bandwidths, the

calculated peak dBm will change according to a 20 log10 B rule.

Most of the UWB devices were measured in the radiated mode at a distance of 1 meter. In the

radiated pulse shape measurements, the pulse shape was not corrected for the effects of the

frequency-dependent receiving antenna gain. However, the spectrum calculated from the pulse

shape via FFTs is corrected for antenna gain and plotted as field strength at 1 meter. The field

strength values are based on peak power in a given computational bandwidth, )f, and plotted in

dBμV/m. The peak field strength can be directly compared to the spectrum analyzer

measurements by using an appropriate bandwidth conversion factor (e.g., 20 log10 B for wider

bandwidths where the UWB pulses are independently resolved). The FFT is based on the
8-4

emission from a single pulse. Therefore, it contains none of the spectrum fine structure caused

by a train of impulses and the associated pulse train modulation techniques.

Major objectives for FFT calculation of emission spectra include a comparison with spectrum

analyzer measurements. Close agreement between FFT and measured spectra ensures

confidence in the accuracy of the full-bandwidth pulse shape measurements and the adequacy of

narrowband spectrum analyzer measurements in determining the RF envelope of UWB

emissions. Since the Device A full bandwidth pulse shape (and the corresponding FFT spectra)

was measured in the conducted mode, while the spectrum analyzer measurements in the

following sections were made in the radiated mode, no direct quantitative comparison of the full

bandwidth measurements and the spectrum analyzer measurements is possible. A full set of

radiated measurements were made with many of the other UWB devices, however, and these

show good quantitative comparison between the full bandwidth data and the spectrum analyzer

data. On a qualitative basis, the Device A conducted FFT spectrum shows the same

concentration of energy in the 5300-6100 MHz region shown in the radiated spectrum analyzer

measurements described next.

8.2.4. Narrowband Peak Emission Spectra

Figure 8.3 contains a series of emission spectra measured at a distance of 1 m with a spectrum

analyzer using a peak detector and bandwidths of 10 kHz to 3 MHz, as described in Section 6.4.2. 

This data is calibrated using two scales: The left-hand scale is in dBm, referenced to the antenna

terminals of an imaginary antenna having a constant aperture equivalent to what a 5.9 dBi

antenna would have at 1 GHz. This calibration removes the effects of a frequency-dependent

receiving antenna aperture (unlike the full-bandwidth pulse measurements), leaving the dBm

value unchanged at 1 GHz. The right-hand scale is in dBμV/m at a distance of 1 m. The details

of these calibrations can be found in Appendix C.

Major objectives for these measurements include development of techniques for UWB spectrum

measurements using commercial off-the-shelf (COTS) equipment, investigation of the relative

utility of -10-dB and -20-dB and mid-band frequency points on the UWB spectra, peak detector

bandwidth correction factors, investigation of preferred spectrum measurement bandwidths for

various regulatory and modeling purposes, and comparison with spectra derived by FFT processing of full bandwidth pulse shapes. Successful comparison between the FFT-derived emission

spectra and spectrum analyzer measurements would strongly suggest that spectrum analyzer

measurements in a narrower bandwidth are accurate and adequate for purposes of describing the

overall UWB emission spectrum, along with the -10-dB and -20-dB frequency points. 

If the measurement system bandwidth is sufficiently greater than the UWB device PRR, the

UWB impulses are resolved in the measurement system IF as independent, non-overlapping

pulses – also called “impulsive” behavior. The peak value of these pulses is expected to depend

on the measurement bandwidth, B, according to a 20 log10 B rule. This would result in

approximate 10-dB differences between successively larger measurement bandwidths, when
8-5

-130

-120

-110

-100

-90

-80

-70

-60

-50

-40

-30

4700 4900 5100 5300 5500 5700 5900 6100 6300

10

20

30

40

50

60

70

80

90

100

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

f2r1-7.ep

Frequency (MHz)

Peak Power Through Constant-Aperture Antenna (5.9 dBi at 1 GHz) @ 1 m (dBm)

Peak Field Strength at 1 m (dBuV/m)

Figure 8.3. Device A measured spectra as a function of bandwidth.

those bandwidths increase in a 1, 3, 10 progression. In this example, the 10-kHz PRR of Device

A causes the peak signal to appear impulsive for all measurement bandwidths greater than 10 kHz

(i.e. all of the bandwidths measured in Figure 8.3). 

If the measurement bandwidth is less than the UWB PRR, the UWB energy is stretched out in

time in the receiver IF such that successive UWB pulses overlap, causing the appearance of a

more-or-less continuous signal. Depending on the timing between successive UWB impulses,

the IF signal can appear like a continuous wave (CW) signal, or Gaussian noise, or other

modulations. Signals that appear noise-like are expected to follow a 10 log10 B rule. This would

result in approximate 5-dB increases between successively larger measurement bandwidths,

when the bandwidths follow a 1, 3, 10 progression. Signals that have a CW appearance do not

change in amplitude as the measurement bandwidth is increased. In this example, since no

emission spectrum measurements were made with bandwidths less than 10 kHz, there was no

opportunity to observe the behavior of signals from Device A in the noise-like or CW region. 

Most of the energy in this spectrum is contained in the 5300-6100 MHz range. These spectra have

numerous irregular lobes that were about 250 MHz wide, with 10-20 dB nulls between them. 

Detailed measurements showing UWB signal behavior as a function of higher spectrum

resolution, bandwidth, and/or time, as described in the remainder of this section, were made near

5700 MHz. The measurements near 5700 MHz show an impulsive behavior (10 dB between

successive measurement bandwidths). Measurements made near 4800 MHz or 6200 MHz tend

to show 5-dB differences between successive bandwidths. This was probably because the UWB

peak signals were not sufficiently above the measurement system noise levels near these
8-6

Figure 8.4. Device A spectrum fine structure (5700 MHz).

frequencies, and the measurements were substantially affected by system noise (5-dB difference

between successive detector bandwidths).

8.2.5. Spectrum Fine Structure

The details of the spectrum fine structure are a result of the techniques used for dithering or

modulating the pulse train. In general, various measurements of the detailed spectrum finestructure were attempted for all UWB devices, but these measurements sometimes provided

inconclusive results that were not included in this collection of significant measurement results. 

The purpose of these measurements was to gain information on PRR and modulation details so

that later measurements of APDs and detector values could be understood more completely. 

Measurements of spectrum fine-structure may reveal unexpected details like the repetition

frequency of the sequence used to dither the impulse train, etc. Specifically, non-dithered pulse

trains will show discrete spectral lines at harmonics of the PRR. When the basic PRR is dithered

with a sequence that repeats at a certain rate, discrete spectral lines will appear separated by that

rate (frequency). The relative amounts of energy in the spectral lines caused by the dither

sequence, the spectral lines caused by the PRR, and the continuous spectral background are

determined by the modulation and dithering details.

Device A (Figure 8.4) showed major spectral lines at a 10-kHz spacing, as expected from the 10-

kHz PRR that was measured in the time domain. A closer examination showed the additional

fine structure of distinct spectral lines on either side of the main lines about 1700 Hz away. The

cause of these sidebands is not known.
8-7

-90

-80

-70

-60

-50

-40

-30

-20

-10

0

10

100 Hz

300 Hz

1 kHz

3 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 M Hz

3 MHz

f2r8.ep

Signal Relative to 3 MHz as a Function of Bandwidth (dB)

Figure 8.5. Device A, peak signal bandwidth progression stairstep .

8.2.6. Bandwidth Progression Stairstep Measurements

This measurement shows the values of peak detector readings made with a range of measurement

bandwidths (100 Hz - 3 MHz). Specifically, it shows that the UWB device PRR causes impulsive

behavior (10-dB steps between successive measurement bandwidths) for bandwidths greater than

the PRR. For bandwidths less than the PRR this graph will show noise-like (5-dB steps between

successive bandwidths), CW-like (constant values independent of bandwidth), and various other

behaviors.

 

Figure 8.5 shows the peak signal level bandwidth progression measurements for Device A (PRR

= 10 kHz), showing 10-dB spacing for bandwidths greater than 10 kHz. The apparent departure

from 5-dB spacing for some of the narrower measurement bandwidths is caused by: 1) the details

of the dithering technique, 2) the measurement frequency, and 3) the lower number of

independent samples (which limits the ability of the peak detector to reach statistically valid peak

values). This last factor is apparently not present in the Device A measurements, but it can be

seen clearly in the measurements of some of the other devices (e.g. Device B, Figures 8.20 and

8.21). It is present in those cases because the rate at which independent samples occur is

proportional to the measurement bandwidth, and the decreased bandwidth results in fewer

independent samples during the fixed time period of these samples. This accounts for the jagged

appearance of the narrowest bandwidths, where many of the time samples failed to contain any

of the statistically rarer peaks that are 8-10 dB above the average values. For these cases, the 5-dB

rule appears to be maintained more closely if one considers the peak value seen within the whole

stair “step.” Using the whole “step” has the effect of increasing the number of independent

samples for the lowest bandwidths, partially correcting the problem of too few samples. 
8-8

0.2

0.4

0.6

0.8

1.0

0

0.2x10

-4

0.4x10

-4

0.6x10

-4

0.8x10

-4

1.0x10

-4

1325 MHz

f3r16.ep

Time (seconds)

Log Volts

Figure 8.6. Device B, pulse train PRR,16 kBit/second mode,

 external detector.

8.2.7. Gating, PRR, and Modulation

These observations are intended to show the occurrence of patterns of pulses, as measured in the

time domain at a single frequency. The purpose of these measurements was to help gain detailed

information on PRR and modulation so that later measurements of APDs and detector values

could be understood more completely. Depending on the features employed in a specific UWB

device, these measurements could show the basic PRR (Figure 8.6), the pattern of gated pulse

bursts (Figure 8.23), or the modulation method used to add dithering or data to the pulse trains

(Figure 8.26). Such modulation methods could include on-off modulation of individual pulses,

fixed or variable pulse delays referenced to an absolute time base, or fixed or variable pulse

delays referenced to the preceding pulse. These measurements can be gathered under a wide

range of bandwidths, time scales, and detector types, as needed to best show the details of the

significant features.

Figure 8.6 shows the appearance of the Device B as a function of time, as seen using a wideband

RF detector with logarithmic compression. This figure shows a 100-kHz PRR gated pulse train,

without apparent dithering or other modulation.

In addition to dithering (used to make UWB emission spectra look more noise-like) and

modulation (employed to provide for transmission of data), some UWB systems also employ

gating. A gated system employs a programmed set of periods where the UWB transmitter is

turned off or on for a period of many UWB pulses. For example, a UWB system might transmit

a gated burst of data lasting 10 ms, followed by a 40-ms period where no pulses are transmitted. 
8-9

These so-called “gated” modes are different from data encoding techniques, where individual

pulses are turned off or on to encode a digital message. 

8.2.8. Amplitude Probability Distribution (APD)

The APD contains information on the percentage of time the envelope of UWB signals in a

specific IF bandwidth exceeds various amplitudes. A more complete description of APDs is

included in Appendix A. Figure 8.7 contains APDs from Device D, 1-MHz PRR, 100% gating

(i.e., transmitting continuously). These APDs were measured in bandwidths between 10 kHz and

20 MHz at a single center frequency near the frequency of maximum UWB spectral power. With

some devices, two sets of APDs were measured to distinguish between different signal fine

structures seen at different frequencies. Some UWB devices measured earlier in the measurement

program did not include the 10-MHz and 20-MHz bandwidth measurements, as this capability

was not yet available to us.

Since the APD will change substantially as a function of measurement bandwidth (or in victim

receivers having various bandwidths), the graph contains a family of APDs measured in

bandwidths between 10 kHz and 3 MHz, and sometimes including10-MHz and 20-MHz

bandwidths. The APD provides a conceptual technique for understanding how various UWB

signals will appear within victim receivers of various bandwidths. In particular, since the raw bit

error rate of a victim receiver is closely related to the probability that the interfering UWB signal

will exceed the amplitude of the desired signal, the APD provides a tool to relate UWB

characteristics to the interference caused to various victim systems. (Note that many modern

systems employ a variety of error correction techniques that are intended to correct most raw bit

errors, so the number of errors presented to the system user may be related to the APD in more

complex ways.)

The APD graph is plotted in power (dBm) versus the percentage of time (or probability) that the

signal envelope will exceed a specified level. The percentage-of-time scale is weighted as 0.5

log10(-ln P(A>a)), as described in Appendix A. This particular weighting function has two

important consequences. First, the envelope of Gaussian noise (including receiver noise) plots as

a straight line, making it easy to recognize when UWB signals act in a non-Gaussian manner. 

Second, these APD graphs never reach 0% or 100%, but only approach these percentages. This

reflects the real world circumstance that the maximum and minimum peak amplitudes of many

phenomena (including Gaussian noise) are dependent on the total number of independent

measurements included in the sample.

In the APD graph, the individual APDs are labeled by IF measurement bandwidth. The amplitude 

(power) scale is in dBm, referred to the output terminals of the measurement antenna, which was

located 1 m from the UWB source. The gain of the ridged-horn measurement antenna was

between 5 dBi and 10 dBi, depending on the measurement frequency. Although the APD

calibration could have been converted to field strength, the primary object of the APD
8-10

-150

-130

-110

-90

-70

-50

.001 .1 1 5 10 20 35 50 70 80 90 95 99

20 MHz

10 MHz

3 MHz

1 MHz

300 kHz

100 kHz

30 kHz

10 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

10 MHz

20 MHz

r_d3_b3.ep

Percent Exceeding Ordinate

Power (dB

m)

Figure 8.7 Device D APDs, 1-MHz PRR, 100% gating.

measurements and analysis was to show the relative behavior of UWB signals as a function of

measurement bandwidth (where absolute field strength calibration was less important). 

Most APDs show two distinct patterns, depending on the relationship of the measurement

bandwidth to the PRR. For measurement bandwidths greater than the PRR (bandwidths of 3 MHz

and above for Device D, PRR = 1 MHz), the UWB signals look impulsive. This means that the

individual UWB RF impulses are converted to independent pulses within the receiver IF

bandwidth. The peak amplitude of these IF pulses varies as 20 log10 B, and the duration of these

pulses is approximately 1/B. For example, in the 10-MHz bandwidth, each pulse will last

approximately 0.1 μs. With a PRR of 1 MHz, the total duration of these pulses adds up to a total

of 0.1 s each second. This is a total pulse duration equal to about 10% of the total time, matching

the approximate 10% “plateau” on the 10-MHz APD in Figure 8.7. 
8-11

The peak amplitude of the plateau is expected to vary according to a 20 log10 B rule for

independent pulses. Thus, the 20-MHz APD should have a peak value 16.5 dB greater than the 3-

MHz APD – very close to the difference in Figure 8.7.

As shown in the graph, the Device D pulse was present in the 20-MHz bandwidth about 5% of the

total time. During the 95% of the time that the Device D pulse was not present in the

measurement receiver, the only “signal” present in the measurement receiver was receiver system

Gaussian noise. Thus, the remainder of the APD (after the impulsive signals have been

subtracted) is only system noise. In the example, the 20-MHz APD tends to match a straight line

drawn on the graph corresponding to Gaussian noise over the percentage ranges between about

20% and 80%. This straight line corresponds to a 20-MHz receiver bandwidth with a noise figure

of 10 dB. The APD of system noise corresponding to other bandwidths and noise figures would

be represented by a straight line with an identical slope, but offset vertically and intercepting the

37% value at Noise (dBm) = NF + 10 log10 B(Hz) -174, where NF is the measurement system

noise figure in dB.

The departure of the measured APDs from the expected Gaussian noise straight line (the 80-99%

range on the 20-MHz APD, and various amounts on other curves) is not completely resolved, but

it is probably due to insufficient video bandwidth in the measurement system that was assembled

to make the 10-MHz and 20-MHz bandwidth measurements. As expected, the departure-fromideal-response was greater for the wider bandwidths.

Within measurement bandwidths equal to or less than the UWB PRR (1-MHz and less, in this

example), the UWB energy does not appear as independent pulses. In this “non-impulsive” mode,

the energy from successive UWB pulses is sufficiently stretched out in time by the IF bandwidth

filter that the “pulses” overlap. Depending on the timing between pulses, the overlapping pulses

can constructively or destructively interfere in the receiver IF bandwidth. This process can

produce various results, including a series of discrete spectral lines at the harmonics of a uniform

undithered PRR, or a continuum of energy resembling Gaussian noise for modulated or dithered

UWB pulse trains. The use of a repeated binary sequence to dither the pulse train can produce

discrete close-spaced spectral lines at frequencies related to the repetition rate for the whole

sequence.

Although some dithering or modulation techniques may produce signals that appear like Gaussian

noise in narrow bandwidths, this apparent Gaussian noise is different in origin from the Gaussian

noise that is present in the wider bandwidth impulsive modes and in gated modes. In the

narrowband mode (bandwidths less than the PRR) the energy from ungated UWB impulses is

continually present in a narrow IF bandwidth, and the measured signal (often a signal with

Gaussian characteristics) comes from the UWB energy. In wideband and gated modes, there are

time intervals where no UWB energy is present, and measurement system (Gaussian) noise

appears whenever the UWB energy is absent. Therefore, although the APDs measured in these

two cases may both appear like Gaussian noise, it is important to note that for gated signals or

wide measurement bandwidths the low amplitude Gaussian portion of the APD is not caused by
8-12

the UWB signal. In the case of the Device D example, the amplitude of the Gaussian noise in the

30-kHz APD is about 17 dB higher than would have been calculated on the basis of system noise

figure and bandwidth.

8.2.9. Detector Summary

While the complete APD data set may give good insights on how specific UWB devices will

interact with specific victim receivers, a specification for computing a single numerical value from

any APD may be desirable for comparing UWB emissions with regulatory limits. APDs can be

processed to give numerical values for detector functions (called “statistics” in Appendix A) like

peak, RMS, average voltage, average logarithm, median, etc. The specific intent of this section is

to explain the numeric values produced by various detectors and measurement bandwidths, based

on the particular characteristics of UWB signals. These results are intended to give information on

which detector functions and bandwidths are most suited for specifying Part 15 limits, rather than

to suggest specific numerical values for these limits. 

Since the APD contains only first-order statistics, presumably any detector function that can be

derived from first-order statistics can be computed from the APD. The detector summary analysis

computes the following detector values from APDs measured in various bandwidths: peak, RMS,

average voltage, and average logarithm. In some EMI receivers and newer spectrum analyzers

these statistics are made available by hardware detectors or real-time digitizer/signal processors

designed to process the IF signal in comparable ways. The quasi-peak detectors specified by the

FCC for most Part 15 measurements below 1 GHz cannot be computed from the APD, because

quasi-peak measurements are a function of second-order statistics, as well as first-order statistics.

8.2.10. FCC Proposed Part 15 Measurement Procedures

The FCC suggested in the UWB NPRM that UWB devices be measured according to the existing

Part 15.35 (b) measurement techniques referred to in Part 15.209. These measurement techniques

include quasi-peak detectors for frequencies below 1 GHz and a pair of measurements (average

and peak) for frequencies above 1 GHz. For frequencies above 1 GHz, the FCC specifies in 15.35

(b) that measurements shall be made using an “average detector function.” It also states that a

maximum peak UWB value should be no more than 20 dB above the average limits, and that

(unless otherwise specified) measurements should be made with a minimum bandwidth of 1

MHz. Based on this text, we believe that the FCC has described a Part 15 average detector

function corresponding to the average voltage value computed from the 1-MHz APD. 

However, the FCC has proposed in their NPRM (Question #50) that Part 15 average

measurements be made with a spectrum analyzer using 1-MHz bandwidth, video filtering

(between 10-kHz and 10-Hz bandwidth) and a peak detector. No guidance is given in the NPRM
2

In a separate communique related to the NTIA UWB test plan, the FCC indicated that the

intention was to employ the spectrum analyzer in the logarithmic IF mode. 

8-13

concerning whether logarithmic or linear IF compression is to be used.2 It is likely that the use of

a logarithmic IF amplifier would give a result very similar to the average logarithm value computed

from the 1-MHz APD. If a linear IF were used instead of a logarithmic IF, the result should be

very similar to the average voltage value computed from the 1-MHz APD. To confirm this

possibility, we also performed an independent measurement that followed the FCC procedures

exactly, using a 1-MHz bandwidth, logarithmic IF mode, video filtering, and sample detector. The

function of the FCC peak detector is duplicated by manually reading the highest value from the

resulting graph. This modified procedure makes it possible to understand the role of sufficient

lowpass video filtering in the suggested FCC measurement. This result is labeled “FCC Part 15"

and represented by a hollow square in the detector summary graphs.

Detector functions corresponding to peak, RMS, average voltage, and average logarithm have

been computed from the APDs measured in each bandwidth and plotted at the corresponding

bandwidth for Device D in Figure 8.8. Details on this computation are included in the discussion

of APDs in Appendix A. These four detector functions tend to emphasize particular parts of the

APD. The peak detector looks only at the single highest measurement value, ignoring all other

measurement values. The RMS detector performs an integration of average power. Because

power is related to “voltage-squared,” the effect of the higher amplitudes in the APD is enhanced. 

The average voltage detector tends to be affected more equally by the whole range of values. The

average logarithm detector gives greatest weight to the lower values. The detectors are normalized

to give the same value for a CW signal. 

Some common “rules-of-thumb” for these detectors (which should be tested against the detector

values actually obtained) include:

1. For Gaussian noise, the peak detector value is typically 8-11 dB greater than the

RMS value (depending on the number of samples), the average voltage is 1 dB less

than the RMS value, and the average logarithm is 2.5 dB less than the RMS value. 

The RMS value is exceeded 37% of the time.

2. For non-overlapping pulses, the peak value varies as 20 log10 B.

3. The RMS value varies as 10 log10 B 

4. Peak $ RMS $ average voltage $ average logarithm.

Figure 8.8 shows the detector summary graph for Device D, 1-MHz PRR, 100% gating. Dashed

lines have been drawn on the graph, corresponding to 10 log10 B and 20 log10 B slopes. These lines

are intended as graphical aids, not as suggested conclusions.
8-14

Figure 8.8. Device D detector summary,1-MHz PRR, 100% gating.

The detector summary shows two major regions: the impulsive region (B >> PRR) and the noiselike region (B >> PRR). In the noise-like region, the values for the RMS, average voltage, and

average log detectors are closely spaced (2.5 dB theoretical difference between average log and

RMS for Gaussian noise). The peak reading is about 8 dB above the RMS (the peak value is

theoretically 8-11 dB above the RMS value for Gaussian noise). The values for all detectors

change according to a 10 log10 B trend. The similarity between the Device D detector readings and

the corresponding readings that would have been theoretically produced by Gaussian noise

suggest that the Device D signal dithering was selected to make it look similar to Gaussian noise.
8-15

A completely different pattern of detector function values is found in the impulsive region (B >>

PRR). There is a much wider divergence between various detector values at any single bandwidth,

and this divergence becomes greater as the bandwidth increases. The peak values follow a 20 log10

B trend line, and the RMS values follow a 10 log10 B trend line. The value of the average voltage

and average log detectors drop substantially when the wider bandwidths allow low amplitudes

associated with measurement system noise to be seen between the duration of UWB impulses. 

Since the average log value emphasizes the lowest amplitude signals, the average logarithm value

is most dramatically affected when the IF output drops to the measurement system noise level

between impulses. The difference between peak values and average logarithm values is as much

as 35 dB for the 20-MHz bandwidth. These relatively large differences between detector values

suggest that the detector functions used in measuring Part 15 regulatory limits must be specified

unambiguously.

The smallest differences between the various detector functions is often seen when B = PRR. In

this example, there is only a 6-dB difference between average logarithm and peak values for the 1-

MHz values.

The FCC-proposed Part 15 measurement procedure using a spectrum analyzer with 1-MHz

bandwidth, logarithmic IF, video filtering, and peak detection was used to make a direct

measurement of the Device D signal. This measurement produced a value of about -84 dBm,

which was essentially identical to the average logarithm detector value computed from the 1-MHz

bandwidth APD. 

In addition to the dBm scale on the detector summary graphs, a level has been marked with a

horizontal dashed line. This level corresponds to the FCC Part 15 limits from 15.209, which is

specified as 500 μV/m at a distance of 3 meters. Since all of the measurements in this section of

the report were measured at a distance of 1 meter, the FCC limit has been adjusted for a 1-m

distance and the gain of the antenna used for the measurement. More details on this conversion

can be found in Appendix C. 

8.3 Measured Data Summaries for Devices

The following sections contain compilations of measured data from UWB devices A-E and an

electric drill. Additional measured data on each device is contained in a corresponding separate

section in Appendix D.
8-16

8.3.1 Summary of Device A Measurements

Device description. Device A uses a 10-kHz PRR, apparently not gated, dithered, or modulated. 

ITS had no information on the operational aspects of this device, and testing proceeded on the

assumption that the on-off switch was the only significant test variable.

Full bandwidth pulse shape. Figure 8.9 shows a complex pulse shape with multiple lobes,

probably caused by a multipole sharp-cutoff RF filter giving a tightly-defined UWB spectrum. 

FFT emission spectrum. Figure 8.10 shows most UWB energy is in the 5100-6100 MHz band.

Narrowband peak emission spectra. Spectrum analyzer measurements (Figure 8.11) show

most energy in the 5100-6100 MHz range. This spectra had 5-6 irregular lobes about 250 MHz

wide, with 10-20 dB nulls between them.

Spectrum fine-structure. Figure 8-12 shows details near 5700 MHz reflecting the 10-kHz PRR

of Device A .

Bandwidth progression stairstep. Figure 8-13 clearly shows 10-dB steps for bandwidths

greater than the 10-kHz PRR (impulsive behavior) and 5-dB steps (noise-like behavior) for

bandwidths less than 10 KHz .

APDs. Figure 8-14 shows impulsive behavior for all measurement bandwidths greater than 10

kHz. However, although the peak values followed a 20 log10 B trend line for the narrower

bandwidths, the 1-MHz and 3-MHz peak values (Figure 8.15) were appreciably lower than the

trend line (8 dB lower at the 3-MHz bandwidth). This 8-dB difference was traced to a deficient

component in the spectrum analyzer, which was found to provide insufficient spectrum analyzer

video bandwidth and was replaced by the manufacturer. Unfortunately, Device A had already

been returned to its owner when the repaired spectrum analyzer became available. It should

probably be assumed that a re-measurement of Device A would have placed the 1-MHz and 3-

MHz readings on the proper trend lines.

Detector summary. We note in Figure 8-15 that the detector values are within 8 dB of each

other in the 10-kHz measurement bandwidth, and they diverge rapidly as the bandwidth is

increased. The difference between peak and average logarithm is as much 40 dB for the 3-MHz

bandwidth (presumably, this difference would have been 8 dB greater – a 48-dB difference – if

the spectrum analyzer had been working properly during these measurements).

FCC Part 15 measurement. The FCC-recommended Part 15 measurement procedure (1-MHz

bandwidth, logarithmic IF, video filtering, and peak detection) produced a value of -112 dBm

with a 300 Hz video bandwidth (300 Hz was the minimum video bandwidth for the spectrum

analyzer when used with 1-MHz resolution bandwidth). This matches the -112 dBm average

logarithm detector value in Figure 8-15 computed from the 1-MHz bandwidth APD. 
Figure 8.12. Device A spectrum fine structure.

-80

-60

-40

-20

0

4000 5000 6000 7000

Frequency (MHz)

dBm

Figure 8.10. Device A, conducted power spectrum,

)f = 16.67 MHz.

-2.5

0

2.5

0 5 10 15 20 25

Time (nanoseconds)

Volts

Figure 8.9. Device A full bandwidth pulse shape.

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-120

-110

-100

-90

-80

-70

-60

-50

-40

-30

4700 4900 5100 5300 5500 5700 5900 6100 6300

10

20

30

40

50

60

70

80

90

100

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

f2r1-7.ep

Frequency (MHz)

Peak Power Through Constant-Aperture Antenna (5.9 dBi at 1 GHz) @ 1 m (dBm)

Peak Field Strength at 1 m (dBuV/m)

Figure 8.11. Device A spectra as a function

of measurement bandwidth.

8-17
8-18

-90

-80

-70

-60

-50

-40

-30

-20

-10

0

10

100 Hz

300 Hz

1 kHz

3 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

f2r8.ep

Signal Relative to 3 MHz as a Function of Bandwidth (dB)

Figure 8.13. Device A peak amplitude as a function of bandwidth.

-150

-130

-110

-90

-70

-50

.001 .1 1 5 10 20 35 50 70 80 90 95 99

3 MHz

1 MHz

300 kHz

100 kHz

30 kHz

10 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

Percent exceeding ordinate

dBm

Figure 8.14. Device A APDs.

Figure 8.15. Device A detector summary.
8-19

8.3.2 Summary of Device B Measurements

Device description. Device B operates in four data rate modes, ranging between 16 kb/s and 128

kb/s. Any of these data rates can be used to support a two-way voice circuit or a two-way data

circuit. ITS measured the device in its 16-kb/s and 128-kb/s voice modes, but did not measure

the intermediate 32-kb/s and 64-kb/s modes. 

The 16-kb/s mode provides an average user data rate of 16 kb/s, using a transmitted signal of

about 100 K impulses per second, which is gated on for 8.8 ms and gated off for 35.2 ms (44 ms

total cycle time). During the 8.8 ms when the signal is gated on, the data stream is modulated on

the train of impulses by deleting selected pulses (on-off keying). Corresponding gating and PRR

values for the other modes can be found in Table 8.1.

Table 8.1. Burst and PRR Timing for Various Data Rates

Data Rate

(kb/s)

Burst length

(on time, ms)

Burst Repetition

time (ms)

Burst duty

cycle

(gating)

Pulse repetition rate (PRR)

16 8.8 44 20% 100 kHz

32 4.4 22 20% 200 kHz

64 2.2 11 20% 400 kHz

128 2.2 5.5 40% 400 kHz

Full bandwidth pulse shape. Device B emission spectrum is tightly filtered to limit it to a 500

MHz-wide bandwidth This filtering produces relatively complex and extended pulse shapes,

lasting up to 30-40 ns (Figure 8.16). 

FFT emission spectrum. The FFT-derived spectrum (Figure 8.17) shows a sharply-defined 500-

MHz RF bandwidth. The peak signal in the FFT-derived spectrum is 116 dBμV/m in a bandwidth, )f, equal to 20 MHz. 

Narrowband peak emission spectra. Figures 8.18 and 8.19 show a sharply-defined bandwidth

between 1250 and 1800 MHz, with 5-6 ripples of 5-10 dB depth across the passband. The peak

signal measured with a spectrum analyzer in a 3-MHz bandwidth (Figures 8.1 or 8.19) is

approximately 100 dBμV/m. Since the UWB signal for Device B is impulsive at bandwidths

greater than the PRR ( maximum PRR = 400 kHz) a 20 log10 B1/B2 correction factor is needed to

convert the 3-MHz bandwidth data to 20-MHz data. This correction factor equals16.5 dB. This

converts the 3-MHz bandwidth value measured by the spectrum analyzer to the corresponding

20-MHz bandwidth value of 116.5 dBμV/m, which compares closely with the FFT-computed

value of 116 dBμV/m.
8-20

Bandwidth progression stairstep. The stair step bandwidth progression figures (Figures 8.20

and 8.21) suggest that Device B produces signals that appear noise-like for measurement

bandwidths less than the 100-kHz or 400-kHz PRR’s associated with the respective 16-kb/s and

128-kb/s modes, made at 1325 MHz and 1500 MHz respectively.

Gating, PRR, and modulation. Figures 8.22 and 8.23 show the 20% and 40% gating duty cycles

for the 16-kb/s and 128-kbs/s modes, respectively. Figures 8.24 and 8.25 show the basic PRRs

for the 16-kb/s and 128-kb/s modes, respectively. Figure 8.26 shows an example of a missing

pulse, caused by the on-off pulse modulation of the Device B data stream.

APDs. Both APDs (Figures 8.28 and 8.29) were measured at 1545 MHz. These APDs were

inadvertently measured using video filtering equal to the resolution bandwidth, which prevented

the low-amplitude portion of the APD from reaching minimum values. This effect is believed to

have prevented this portion of the APDs from following the expected straight line. This problem

was recognized and corrected in subsequent APD measurements for other devices. The very

steep sides on the “plateau” at 20% and 40% respectively for the 16-kb/s and 128-kb/s modes

reflect the respective 20% and 40% gating modes.

Detector summaries. The detector summaries (Figures 8.30 and 8.31) show approximately a 30-

dB difference between peak and average logarithm and a 20-dB difference between average

logarithm and RMS values for the narrower bandwidths, caused mainly by the gating modes.

During the gated-off period, system noise is the only signal contributing to the APD. Since the

average logarithm value is affected most by low amplitude, high-percentage signals, measurement

system noise has a strong effect on the average logarithm value of gated signals. 

The peak value follows a 10 log10 B trend line (like Gaussian noise) for bandwidths less than the

PRR and a 20 log10 B trend (like an impulse) for bandwidths greater than the PRR. Peak

measurements are identical for both 3-MHz bandwidth APDs, where impulses are non-overlapping. For the lower bandwidths where the signal acts like Gaussian noise, peak measurements for

each PRR differ by about 6 dB, which would be expected from the 4:1 ratio in the number of

pulses/s (not counting the percentage gating ratio).

The RMS values follow a 10 log10 B line for all bandwidths. The best-fit trend lines for the two

modes are about 8 dB apart. A difference of 9 dB would have been expected, based on the 8:1

ratio in the total number of impulses (including the effect of the 2:1 change in gating ratio).

FCC Part 15 measurement. A measurement similar to the Part 15 measurement (except using a

3-MHz bandwidth) was done at 1650 MHz, 128 kb/s mode with logarithmic IF, 10-Hz video

bandwidth, and sample detection (Figure 8.27). This gave an maximum measurement of about -

74.5 dBm, with less than 1-dB time ripple caused by the 5.5 ms gating cycle. Compensating for

the gain of the 19 dB preamplifier, the “Part 15-like" value was -93.5 dBm, which is within 2 dB

of the average logarithm computed from the 3-MHz APD. No Part 15 measurements were made

at 1-MHz bandwidth or using the 16-kb/s mode.
-120

-110

-100

-90

-80

-70

-60

-50

-40

-30

-20

1150 1250 1350 1450 1550 1650 1750 1850

20

30

40

50

60

70

80

90

100

110

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

f2r1-6.ep

Frequency (MHz)

Peak Power Through Constant-Aperture Antenna (5.9 dBi at 1 GHz) @ 1 m (dBm)

Peak Field Strength at 1 meter (dBuV/m)

Figure 8-19. Device B, spectra as a function of bandwidth,

128-kb/s voice mode.

70

80

90

100

110

120

1250 1500 1750

Frequency (MHz)

Field Strength (dBuv/m)

Figure 8.17. Device B, radiated peak field strength

at 1 m, )f = 20 MHz.

-0.2

-0.1

0

0.1

0.2

0 10 20 30 40

Time (nanoseconds)

Volts Figure 8.16. Device B full bandwidth pulse shape.

-120

-110

-100

-90

-80

-70

-60

-50

-40

-30

-20

1150 1250 1350 1450 1550 1650 1750 1850

20

30

40

50

60

70

80

90

100

110

f3r1-6.ep

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

Frequency (MHz)

Peak Power Through Constant Aperture Antenna (5.9 dBi at 1 GHz) @ 1 m (dBm)

Peak Field Strength at 1 m (dBuV/m)

Figure 8.18. Device B, spectra as a function of bandwidth,

16-kb/s voice mode.

8-21
-90

-80

-70

-60

-50

-40

-30

-20

-10

0

10

0 20 40 60

f2r7.ep

100 Hz

300 Hz

1 kHz

3 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

Amplitude as a Function of Bandwidth Relative to 3 MHz

Figure 8.21. Device B, 128-kb/s voice mode,

 bandwidth progression stairstep.

-90

-80

-70

-60

-50

-40

-30

-20

-10

0

10

0 20 40 60

f3r25.ep

100 Hz

300 Hz

1 kHz

3 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

Dimensionless units

Amplitude as a Function of Bandwidth Relative to 3 MHz

Figure 8.20. Device B, 16-kb/s voice mode,

bandwidth progression stairstep.

-110

-100

-90

-80

-70

-60

-50

-40

-30

-20

-10

0 0.005 0.010 0.015 0.020

peak detected

1325 MHz

f13r20.ep

Time (seconds)

Received Signal Level (dBm)

Figure 8.23. Device B, 128-kb/s mode,

 3-MHz IF bandwidth.

-90

-80

-70

-60

-50

-40

-30

-20

-10

0

10

0 0.1 0.2 0.3 0.4 0.5

1325 MHz

f3r13.ep

Time (seconds)

Received Signal Level (dBm)

Figure 8.22. Device B, 16-kb/s mode, 3-MHz IF

bandwidth, positive peak detector.

8-22
0.2

0.4

0.6

0.8

1.0

0

0.2x10

-4

0.4x10

-4

0.6x10

-4

0.8x10

-4

1.0x10

-4

1325 MHz

f3r16.ep

Time (seconds)

Log Volts

Figure 8.24. Device B PRR waveform,

16-kb/s mode, external detector.

-0.005

0

0.005

0.010

0.015

0

0.2x10

-5

0.4x10

-5

0.6x10

-5

0.8x10

-5

1.0x10

-5

Missing pulse

at this location

is part of data

encoding scheme

for this device

f13r15.ep

Time (seconds)

Log Volts

Figure 8.26. Device B, waveform showing on-off

modulation, 128-kb/s mode.

-110

-100

-90

-80

-70

-60

-50

-40

-30

-20

-10

0 0.005 0.010 0.015 0.020

sample detector

f13r47.ep 1660 MHz

Time (seconds)

Received Signal Level (dBm)

Figure 8.27. Device B, 128-kBit/second mode,

3 MHz BW, 10-Hz video (Part 15-like).

-0.005

0

0.005

0.010

0.015

0

0.2x10

-5

0.4x10

-5

0.6x10

-5

0.8x10

-5

1.0x10

-5

f13r14.ep

Time (seconds)

Log Volts

Figure 8.25. Device B PRR waveform,

128-kb/s mode, external detector.

 8-23
8-24

-1 30

-1 10

-90

-70

-50

-30

.001 .1 1 5 10 20 35 50 70 80 90 95 99

3 M Hz

1 M Hz

300 kHz

100 kHz

30 kHz

10 kHz

10 kH z

30 kHz

100 kHz

300 kHz

1 M Hz

3 MHz

Percent exce eding ordinate

dBm Figure 8.28. Device B APDs, 16 kb/s.

Figure 8.30. Device B detector

summary, 16kb/s. Figure 8.31. Device B detector summary, 128 kb/s.

-130

-110

-9 0

-7 0

-5 0

-3 0

.001 .1 1 5 10 20 35 50 70 80 90 95 9 9

3 M Hz

1 M Hz

300 kH z

100 kH z

3 0 kHz

1 0 kHz

10 kH z

30 kH z

100 kHz

300 kHz

1 M Hz

3 M Hz

Perc ent exceeding ordin ate

dBm Figure 8.29. Device B APDs, 128 kb/s.

 
8-25

Figure 8.32. Device C dither at starting point.

Figure 8.33. Device C dither 20 pulses after zero point.

8.3.3 Summary of Device C Measurements

Device description. Device C transmits an ungated 2-MHz PRR pulse train over the 1-5 GHz

range. Device C uses a “relative” dithering technique. This dithering technique uses only a small

amount of pulse delay (in this case, 1.25% delay out of 500 ns average pulse spacing), but timing

is referred to the preceding pulse instead of being referred to a fixed time base. Figures 8.32 to

8.35 show how a continually larger timing variation builds up over elapsed time, until the 1.25%

delay (494-500 ns) for individual pulse-to-pulse spacing covers the entire pulse-to-pulse interval.
8-26

Figure 8.34. Device C dither 44 pulses after zero point

Figure 8.35. Device C dither 90 pulses after zero point

Full-bandwidth pulse shape. Figure 8.36 shows a complex pulse with extended ringing.

FFT emission spectra. Figure 8.37 shows a well-defined emission about 600 MHz wide and

centered at about 1800 MHz. The calculated peak radiated field strength is approximately 86

dBμV/m in a 20-MHz bandwidth..

Narrowband peak emission spectra. Figure 8.38 shows emission spectra with a peak about

700 MHz wide, centered at about 1800 MHz, though the details appear somewhat different from

the FFT spectrum. The 20-MHz bandwidth measurements show a measured peak field strength

of 87 dBμV/m, closely matching the FFT spectrum.

Spectrum fine structure. Figure 8.39 shows a subtle set of fine structures related to the 2-MHz

average PRR. The relative dithering technique appears to be effective in eliminating 2-MHz

discrete spectral lines that extend above the surrounding spectrum. The signals at all frequencies
8-27

appear to have about the same maximum power, with a mix of amplitudes that are distributed

across a range between the maximum and as much as 30 dB below the maximum. However,

there is a pattern of higher-than-average-power frequencies repeated every 2 MHz, where the

average signal power is maximized by spending less time at the lower amplitudes. These higherthan-average-power frequencies were designated “on-spectral-lines” frequencies, and intermediate frequencies with less average power were designated “between-spectral-lines” frequencies. 

Bandwidth progression stairsteps. Figure 8.40 shows a bandwidth progression measurement

made at an “on-spectral-lines” frequency. This measurement shows impulsive behavior at 3-

MHz bandwidth, CW-like results for bandwidths between 100 kHz and 1 MHz, and noise-like

behavior for narrower bandwidths.

Gating, PRR, and modulation. Figure 8-41 shows a measurement of the Device C pulse train,

which exhibits no obvious pulse position modulation. Figures 8-42 and 8-43 show the modulated IF waveforms at 1-MHz and 100-kHz bandwidths. Surprisingly, the 100-kHz waveform

contains much more amplitude modulation than the 1-MHz waveform.

APDs. APDs were measured at between-spectral-lines and on-spectral-lines frequencies – 

Figures 8-44 and 8-45, respectively. Measurements for the between-spectral-lines case were also

made using 10-MHz and 20-MHz bandwidths. 

Detector summaries. The two APDs and the corresponding detector summaries (Figures 8-46

and 8-47) show some interesting results. The two dashed lines on each detector summary are

drawn as best-fit trend lines for the RMS detectors for the on-spectral-lines and between-spectrallines measurements, showing about 2.5-dB difference between the two cases. The 3-MHz onspectral-lines RMS value drops to the between-spectral-lines trend line, showing that the effect of

the relatively narrow bandwidth concentration of on-spectral-lines energy disappears when the

measurement bandwidth is widened beyond the 2-MHz fine structure.

The between-spectral-lines detector summary shows a large divergence between different

detector values at 100-kHz bandwidth, including a 15-dB difference between average logarithm

and RMS and a 27-dB difference between average logarithm and peak. The minimum difference

between detector values occurs at 1-MHz bandwidth, where there is only a 6-dB difference

between average logarithm and peak values. These results are particularly interesting (and

somewhat counterintuitive) because they illustrate a UWB modulation technique that produces

impulse-like results (i.e., wide divergence between detector values) at lower bandwidths and CWlike results at higher bandwidths. The proposed FCC Part 15 limits, however, assume that a

UWB signal will appear more CW-like at lower bandwidths (average limit specified in 1-MHz

bandwidth), and more impulsive at higher bandwidths (peak limit specified in B = 50 MHz).

FCC Part 15 measurements. Part 15 measurements were made at the “on-line” frequency using

a 300-Hz video filter, which produced a signal with about 0.5 dB peak-to-peak ripple and a peak

amplitude of -74 dBm. This compares with a computed average logarithm of -74.2 dBm.
40

50

60

70

80

90

1000 2000 3000 4000

Frequency (MHz)

Field Strength (dBuv/m)

Figure 8.37. Device C, radiated peak field strength

at 1 m, )f = 20 MHz.

-0.02

-0.01

0

0.01

0.02

5 10 15

Time (nanoseconds)

Volts

Figure 8.36. Device C full bandwidth pulse shape. 

-120

-110

-100

-90

-80

-70

-60

-50

-40

-30

-20

1000 2000 3000 4000 5000

20

30

40

50

60

70

80

90

100

110

intrson2.ep

Nominal 500 ns (2 MHz) pulse rate,

with 6 ns pulse-to-pulse dither.

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz 3 MHz

10 MHz

20 MHz

Frequency, MHz

Peak Power Through Constant-Aperture Antenna (5.9 dBi at 1 GHz) @ 1 m (dBm)

Peak Field Strength at 1 m (dBuV/m)

Figure 8.38. Device C spectra in multiple bandwidths.

-110

-100

-90

-80

-70

-60

-50

-40

-30

-20

-10

2107 2109 2111 2113 2115

f3r1.ep

Frequency (MHz)

Recevied Signal Level (dBm) Figure 8.39. Device C, peak-detected to show spectral

line patterns in noise.

8-28
0

0.0025

0.0050

0.0075

0.0100

0

0.2x10

-5

0.4x10

-5

0.6x10

-5

0.8x10

-5

1.0x10

-5

f21r6.ep

Time (seconds)

Received Signal Level (log volts)

Figure 8.41. Device C, pulse train waveform for 10

microseconds, external detector.

-130

-120

-110

-100

-90

-80

-70

-60

-50

-40

-30

0 0.01 0.02 0.03 0.04 0.05

f4r4.ep

Sample detection

Time (seconds)

Received Signal Level (dBm)

Figure 8.43. Device C, time waveform in 55 ms,

100-kHz IF bandwidth.

-120

-100

-80

-60

-40

-20

0 0.01 0.02 0.03 0.04 0.05

f4r1.ep

Sample detection

Time (seconds)

Received Signal Level (dBm)

Figure 8-42. Device C, time waveform in 55 ms,

3-MHz IF bandwidth.

-90

-80

-70

-60

-50

-40

-30

-20

-10

0

10

f2r7.ep

100 Hz

300 Hz

1 kHz

3 kHz

10 kHz

30 kHz

100 kHz

1 MHz 300 kHz

3 MHz

Received Power, Relative to 3 MHz Bandwidth (dB)

Figure 8.40. Device C, peak bandwidth

 progression stairstep.

 

 

 8-29
8-30

-140

-120

-100

-80

-60

-40

.001 .1 1 5 10 20 35 50 70 80 90 95 99

20 MHz

10 MHz

3 MHz

1 MHz

300 kHz

100 kHz

30 kHz

10 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

10 MHz

20 MHz

Percent Exceeding Ordinate

Power (dBm)

Figure 8.44. Device C APDs,

between spectral lines.

Figure 8.47. Device C, detector summary

 (on spectral lines)

Figure 8.46. Device C, detector summary

(between spectral lines).

-150

-130

-110

-90

-70

-50

.001 .1 1 5 10 20 35 50 70 80 90 95 99

3 M Hz

1 M Hz

300 kHz

100 kHz

30 kHz

10 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

Percent Exceeding Ordinate

Power (dBm)

Figure 8.45. Device C APDs,

 on spectral lines.

 
8-31

8.3.4 Summary of Device D Measurements

Device Description: Device D operates with basic PRRs of 1 MHz, 5 MHz, and 10 MHz, with

gating duty cycles available between 25% and 100% (continuously on). A fixed 25% dither is

applied, using an absolute time base in all modes. The device was measured in four modes: 1-

MHz PRR, 25% gated duty cycle; 1-MHz PRR, 100% gated duty cycle; 10-MHz PRR, 25%

gated duty cycle; and 10-MHz PRR, 100% gated duty cycle. The gating cycle for both PRRs

utilized a 2-ms “on” period and a 6-ms “off” period. This data transmission device was measured

without transmitting any actual data.

Full-bandwidth pulse shape and FFT spectrum. The Device D radiated full-bandwidth pulse

shape (Figure 8.48) was used to compute an FFT peak spectrum over the 1-4 GHz range (Figure

8.49). 

Narrowband peak emission spectra. Figures 8.50-8.53 contain spectrum measured for each of

four Device D operating modes. These spectra compared closely in shape and amplitude with the

FFT spectrum. Figure 8.53 included measurements made in a 20-MHz bandwidth, which show

close agreement (within 1-2 dB) with the FFT values computed in a 20-MHz bandwidth.

Bandwidth progression stairstep. The stair-step bandwidth progression graphs (Figures 8.54-

8.57) were measured at a frequency near 1560 MHz and show substantial differences between the

1-MHz PRR and the 10-MHz PRR modes. These differences are likely caused by the higher

power and greater frequency separation of the discrete spectral lines in the 10-MHz PRR modes. 

Spectrum fine structure. Details of the spectrum fine structure showed strong lines with1-kHz

spacing in the 1-MHz PRR mode (Figure 8.58) and 10-kHz spacing in the 10-MHz PRR mode

(Figure 8.59). It is not known whether these lines would be present with transmission of actual

data. Figure 8-60 shows pulse-position modulation, measured in a 10-MHz bandwidth.

APDs. The APDs are shown in Figures 8.62 to 8.65. The non-gated modes (100% gating) appear

like Gaussian noise for bandwidths less than the PRR and appear impulse-like for bandwidths

larger than the PRR. The gated modes show clear break-points at the 25% point, with measurement system noise present for the remaining 75% of the duration.

Detector summary. The detector summaries (Figures 8.66 to 8.69) show that the ungated modes

appeared similar to Gaussian noise for bandwidths less than the PRR. For all narrower

bandwidths, peak values were about 10-dB greater than the RMS, and the average voltage and

average logarithm were tightly clustered within a few dB less than the RMS. All of the detector

functions followed a 10 log10 B trend line. The corresponding trend lines were 8-10 dB higher 

for the 10-MHz PRR than for the 1-MHz PRR mode. Only when the bandwidth was greater than

the PRR, did the detector values at a bandwidth begin to diverge. The peak values then followed

a 20 log10 B trend. The average logarithm and average voltage decreased substantially at larger 

bandwidths, when pulses did not overlap and system noise was momentarily the only signal. 
8-32

-0.030

-0.015

0

0.015

0.030

10 20 30

Time (nanoseconds)

Volts Figure 8.48. Device D radiated full-bandwidth

pulse shape

40

50

60

70

80

90

1000 2000 3000 4000

Frequency (M Hz)

Field Strength (dBuv/m)

Figure 8.49. Device D, radiated peak 

field strength at 1 m, )f = 20 MHz.

The gated modes provided peak values identical to the corresponding ungated (i.e.,100%) modes. 

The gated RMS values were approximately 6 dB lower than those of the corresponding ungated

modes, as expected when the total number of impulses was decreased by 75%. The gated mode

average logarithm dropped to within a few dB of system noise (present 75% of the time), with the

10-MHz PRR average logarithm being 1-2 dB higher than the 1-MHz logarithm. 

FCC Part 15 Measurements. Part 15 measurements for the ungated 1-MHz PRR and 10-MHz

PRR modes matched the average logarithms at -84 dBm and -74 dBm, respectively. Part 15

measurements for the gated modes were -100 dBm and -97 dBm for the 1-MHz PRR mode and

10-MHz PRR mode (Figure 8.61), respectively. The 1-2 dB discrepancy between Part 15 and

average logarithm for gated modes was caused by insufficient video filtering of the 8 ms gating

cycle, which allowed 2-3 dB of ripple on the video filter output.

 
-120

-110

-100

-90

-80

-70

-60

-50

-40

1000 2000 3000 4000 5000

20

30

40

50

60

70

80

90

ut13spec.ep

y=y+(13.3*(log(x/1000)))

20 MHz

10 MHz

3 MHz

1 MHz

300 kHz

100 kHz

30 kHz

10 kHz

Cellular

signals

Frequency (MHz)

Peak Power, Through Constant-Aperture Antenna (5.9 dBi at 1 GHz) @ 1 m (dBm)

Peak Field Strength, dBuV/m, at 1 m

Figure 8.53. Device D spectra, 10-MHz PRR, 100% gating.

-120

-110

-100

-90

-80

-70

-60

-50

-40

1000 2000 3000 4000 5000

20

30

40

50

60

70

80

90

ut1spec.ep

3 MHz

1 MHz

300 kHz

100 kHz

30 kHz

10 kHz

Cellular

signals

Frequency (MHz)

Peak Power, Through Constant-Aperture Antenna (5.9 dBi at 1 GHz) @ 1 m (dBm)

Peak Field Strength, dBuV/m, at 1 m

Figure 8.51. Device D spectra, 1-MHz PRR, 100% gating.

-120

-110

-100

-90

-80

-70

-60

-50

-40

1000 2000 3000 4000 5000

20

30

40

50

60

70

80

90

ut17spec.ep

3 MHz

1 MHz

300 kHz

100 kHz

30 kHz

10 kHz

Cellular

signals

Frequency (MHz)

Peak Power, Through Constant-Aperture Antenna (5.9 dBi at 1 GHz) @ 1 m (dBm)

Peak Field Strength, dBuV/m, at 1

Figure 8.52. Device D spectra, 10-MHz PRR, 25% gating.

-120

-110

-100

-90

-80

-70

-60

-50

-40

1000 2000 3000 4000 5000

20

30

40

50

60

70

80

90

ut5spec.ep

3 MHz

1 MHz

300 kHz

100 kHz

30 kHz

10 kHz

Cellular

signals

Frequency (MHz)

Peak Power, Through Constant-Aperture Antenna (5.9 dBi at 1 GHz) @ 1 m (dBm)

Peak Field Strength, dBuV/m, at 1 m

Figure 8.50. Device D spectra,1-MHz PRR, 25% gating.

8-33
-80

-70

-60

-50

-40

-30

-20

-10

0

ut13prog.ep

100 Hz

300 Hz

1 kHz

3 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 Mhz

Level Relative to 3 MHz Bandwidth (dB)

Figure 8.57. Device D bandwidth progression stairstep,

10-MHz PRR, 100% gating.

-80

-70

-60

-50

-40

-30

-20

-10

0

ut17prog.ep

3 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

1 Hz

300 Hz

100 Hz

Level Relative to 3 MHz Bandwidth (dB)

Figure 8.56. Device D bandwidth progression stairstep,

10-MHz PRR, 25% gating.

-80

-70

-60

-50

-40

-30

-20

-10

0

ut1prog.ep

1 kHz

3 kHz

10 kHz

30 kHz

100 kHz

300 kHz

300 Hz

100 Hz

1 MHz

3 Mhz

Level Relative to 3 MHz Bandwidth (dB)

Figure 8.55. Device D bandwidth progression stairstep,

1-MHz PRR, 100% gating.

-80

-70

-60

-50

-40

-30

-20

-10

0

ut5prog.ep

100 Hz

300 Hz

1 kHz

3 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

Level Relative to 3 MHz Bandwith (dB)

Figure 8.54. Device D bandwidth progression stairstep,

 1-MHz PRR, 25% gating.

 

 8-34
-80

-40

0 0.02 0.04 0.06 0.08 0.10

ut17pt15.ep

Time (seconds)

Part 15 Level, in 50 ohms (dBm)

Figure 8.61. Device D, 10-MHz PRR, 25% gating, Part 15

measurement showing ripple in 10-Hz video BW.

0.6

0.7

0.8

0.9

0

0.2x10

-6

0.4x10

-6

0.6x10

-6

0.8x10

-6

1.0x10

-6

10mhz.ep

Time (seconds)

Received Signal Level (dBm)

Figure 8.60. Device D waveforms, 10-MHz IF,

10-MHz PRR, 100% gating, 25% dither, 

showing pulse position modulation.

-120

-100

-80

-60

-40

-20

1559.980 1559.995 1560.010 1560.025

ut5-line.ep

Code length

gives 1 kHz

line spacing

Frequency (MHz)

Measured Power, in 50 ohms (dBm)

Figure 8.58. Device D 1-kHz emission lines,

1-MHz PRR, 25% gating.

-120

-100

-80

-60

-40

-20

1559.90 1559.95 1560.00 1560.05 1560.10

ut13line.ep

Code length

gives 10 kHz

line spacing

Frequency (MHz)

Measured Power, in 50 ohms (dBm)

Figure 8.59. Device D 10-kHz emission lines,

10-MHz PRR, 100% gating.

8-35
8-36

-15 0

-13 0

-11 0

-9 0

-7 0

-5 0

.0 0 1 .1 1 5 10 20 3 5 50 7 0 80 9 0 9 5 99

20 M H z

10 M H z

3 M Hz

1 M Hz

30 0 kH z

10 0 kH z

30 kHz

10 kHz

1 0 kHz

30 kHz

1 0 0 kHz

30 0 kH z

1 M Hz

3 M Hz

10 M H z

20 M H z

r_d3 _b 3.ep

Pe rc en t Exc eedin g Ordinate

Power (dBm)

Figure 8.63. Device D APDs,

1-MHz PRR, 100% gating.

-1 50

-1 30

-1 10

-90

-70

-50

.00 1 .1 1 5 1 0 2 0 3 5 50 70 8 0 9 0 9 5 99

2 0 M Hz

1 0 M Hz

3 M H z

1 M H z

3 00 kHz

1 00 kHz

3 0 kH z

1 0 kH z

1 0 kH z

3 0 kHz

10 0 kHz

1 M Hz

3 00 kH z

3 M Hz

1 0 M H z

2 0 M Hz

r_d1_b3 .ep

Percent Exc eeding Ord ina te

Power (dBm)

Figure 8.65. Device D APDs,

10-MHz PRR, 100% gating.

-15 0

-13 0

-11 0

-9 0

-7 0

-5 0

.0 0 1 .1 1 5 10 20 3 5 50 7 0 80 9 0 9 5 99

20 M H z

10 M H z

3 M Hz

1 M Hz

30 0 kH z

10 0 kH z

30 kHz

10 kHz

10 kHz

3 0 kHz

1 00 kHz

3 0 0 kHz

1 M Hz

3 M H z

1 0 M Hz 2 0 M Hz

r_d4 _d 3.ep

Pe rc en t Exc eedin g Ordinate

Power (dBm)

Figure 8-62. Device D APDs,

1-MHz PRR, 25% gating.

-15 0

-13 0

-11 0

-9 0

-7 0

-5 0

.00 1 .1 1 5 1 0 20 35 50 70 8 0 9 0 9 5 9 9

20 M Hz

10 M Hz

3 M Hz

1 M Hz

30 0 kHz

10 0 kHz

30 kH z

10 kH z

10 kHz

30 kH z

100 kH z

3 00 kH z

1 MH z

3 M Hz

1 0 M H z

2 0 M H z

r_d2 _b3 .ep

Pe rcent Exce eding Ord inate

Power (dBm)

Figure 8.64. Device D APDs,

10-MHz PRR, 25% gating.
8-37

Figure 8.66. Device D detector summary,

1-MHz PRR, 25% gating.

Figure 8.67. Device D detector summary,

1-MHz PRR, 100% gating.

Figure 8.68. Device D detector summary,

10-MHz PRR, 25% gating.

Figure 8.69. Device D detector summary,

10-MHz PRR, 100% gating.
8-38

8.3.5 Summary of Device E Measurements

Device description: Device E is designed to radiate mainly towards the ground. Interchangeable RF heads permitted operation at nominal frequencies of 300 MHz, 900 MHz, and 1500

MHz.. Device E is capable of operating with multiple PRRs.

Full-bandwidth pulse shapes. Full-bandwidth pulse shape measurements were made with all

three RF heads (Figures 8.69, 8.71, and 8.73), with the Device E antennas aimed directly at the

measurement antenna 1 m away. These pulse shapes are very simple, with no additional lobes or

extended ringing.

FFT emission spectra. FFT spectra were computed from each of the measured pulse shapes

(Figures 8.70, 8.72, and 8.74, respectively). 

Narrowband peak emission spectra. Spectrum analyzer measurements were made only with

the 1500-MHz head, with Device E resting on a concrete floor, aiming downwards, with the

measurement antenna 1 meter away from the edge of Device E and slightly above the floor. The

spectrum analyzer measurements signal amplitudes cannot be compared with the FFT spectra

because of different measurement conditions.

Bandwidth progression stairstep. Figure 8.76 shows the Device E emission spectrum, using

the 1500- MHz RF head..

Gating, PRR, and modulation. The mode that was measured included a PRR near 85 kHz and

a gating duty cycle of approximately 25 ms on, 6 ms off (Figures 8-78 and 8-79).A distinguishing

feature of the Device E spectrum analyzer measurements was relatively low received power,

which prevented Part 15 measurements from being made, and which also made it difficult to see

some details of the modulation. 

APDs. Figure 8.80 show APD measurements for the Device E 1500-MHz RF head.

Detector summary. Figure 8.81 shows a detector summary for Device E. Detector values for

bandwidths less than 1 MHz seem to be mostly measurement system noise.

FCC Part 15 measurements. The signal from Device E was apparently below measurement

system noise and Part 15 measurements could not be performed.
-0.5

0

0.5

1.0

0 5 10 15 20

Time (nanoseconds)

Volts

Figure 8.70. Device E full bandwidth pulse shape,

300-MHz RF head, main beam.

70

90

110

130

150

500 1000 1500 2000

Frequency (MHz)

Field Strength (dBuv/m)

Figure 8.71. Device E, (300-MHz head), peak field

strength at 1 m, main beam, )f = 50 MHz.

-0.1

0

0.1

4 8 12

Time (nanoseconds)

Volts

Figure 8.72. Device E full bandwidth pulse shape,

900-MHz RF head, main beam.

85

95

105

115

1000 2000 3000 4000

Frequency (MHz)

Field Strength (dBuv/m)

Figure 8.73. Device E (900-MHz head) peak field

strength at 1 m, main beam, )f = 50 MHz.

8-39
75

85

95

105

1000 2000 3000 4000

Frequency (MHz)

Field Strength (dBuv/m) Figure 8.75. Device E (1500-MHz head) peak field

strength at 1 m, main beam, )f = 50 MHz.

-120

-110

-100

-90

-80

-70

-60

1000 1400 1800 2200

20

30

40

50

60

70

e_spec.ep

Measurement system

noise floor

Frequency (MHz)

Peak Power Through Constant-Aperture Antenna (5.9 dBi at 1 GHz) @ 1 m (dBm)

Peak Field Strength at 1 m (dBuV/m)

Figure 8.76. Device E 1500-MHz RF head, spectra

 as a function of measurement bandwidth.

-60

-50

-40

-30

-20

-10

0

0 10 20 30 40 50 60

e-band.ep

100 Hz

300 Hz

1 kHz

3 kHz

10 kHz

30 kHz

100 kHz

300 kHz

1 MHz

3 MHz

Dimensionless Units

Power Relative to 3 MHz IF Bandwidth (dB)

Figure 8.77. Device E 1500-MHz RF head, peak

bandwidth progression stairstep.

-0.10

-0.05

0

0.05

5 10

Time (nanoseconds)

Volts Figure 8.74. Device E full bandwidth pulse shape,

1500-MHz RF head, main beam.

 8-40
8-41

Figure 8.81. Device E, detector summary. -150

-130

-110

-90

-70

-50

.001 .1 1 5 10 20 35 50 70 80 90 95 99

3 M Hz

1 M Hz

300 kHz

100 kHz

30 kHz

10 kHz

3 MHz

1 MHz

300 kHz

100 kHz

30 kHz

10 kHz

Percent exceeding ordinate

dBm

Figure 8.80. Device E APDs.

-80-70-60-50-40

0 0.05 0.10 0.15 0.20 0.25

e-gating.ep

Time (seconds)

) mBd( ht di wdna Bz HM3 ni re woP

Figure 8.78. Device E pulse train.

-0.000

05

0.00050.0010

0

1x10

-5

2x10

-5

3x10

-5

4x10

-5

5x10

-5

e-train.ep

Time(seconds)

) Bd( stl oVgoL

Figure 8.79. Device E gating waveform.

 
8-42

8.3.6 Summary of Electric Drill Measurements

An electric drill is included in the category of Part 15 Incidental Radiators, since it radiates radio

frequency energy, although its function does not require the generation of radio frequency

signals. The electric drill controls included only an on/off switch (i.e., no electronic speed

control), and it was measured without any mechanical loading on the drill shaft.

Full-bandwidth pulse shapes. The random nature of the timing and amplitude of the electrical

impulses from the drill made it impossible to use a sampling oscilloscope to measure the emitted

impulses. Therefore, no full-bandwidth pulse shape measurements were made, and no FFT

spectrum could be derived from the pulse. 

Narrowband peak emission spectra. The energy radiated from the electric drill was measured

by the spectrum analyzer over the 0.6-5.0 GHz range (Figure 8.82), including a noticeable

“bulge” in the 3.1-4.2 GHz range. The highest frequency of measurable signal is not known. The

apparent change of amplitudes measured with various bandwidths using the peak detector seems

to be about 42 dB for a 2.5-decade change in bandwidth, i.e., a 17 log10 B trend line.

Gating, PRR, and modulation. Figure 8.83 shows the gradually building and declining burst of

noise that occurs about every 17 ms and is related to the 60 Hz power line frequency.

APDs. Each drill APD in Figure 8.84 appears to be divided into impulsive and noise-like parts,

with the noise-like portion determined by measurement system noise and following a 10 log10 B

rule. The impulsive portion, unlike typical UWB devices, assumes a steep, straight-line slope

without any hint of a plateau. The 1-MHz and 3-MHz APDs seem to develop an even steeper

slope at the low-probability end, beginning at the 0.1% point. The APDs were only measured to

.001% (100 k samples); it is likely that the peak values for the wider bandwidths might have been

considerably higher (perhaps 10 dB higher?) had they been extended to the left-hand edge of the

graph. The drill APDs are somewhat problematic, since the peak amplitudes of apparently

independent pulses do not increase as rapidly as 20 log10 B. In addition, the very steep slope of

the low-probability end makes the measured peak very dependent on the sample size.

Detector summary. The drill detector summary (Figure 8.85) shows that the relative values of

the various detectors remain almost the same for all bandwidths; all detectors follow a 10 log10 B

rule. This behavior fails to match the RF spectrum graphs, which show the peak dB values

increasing with a 17 log10 B trend line. This discrepancy can be partly explained because the

peak measurements for the wider bandwidth spectrum measurements contained a much larger

number of independent samples than the narrower bandwidths, which substantially increased the

wider bandwidth peak values. It is also likely that the number of “pulses” seen in the drill

emission increases proportionately with measurement bandwidth–a much different statistical

process than a typical UWB signal (which has a constant number of pulses for wider

bandwidths). The Part 15 measurements were too close to system noise to be useful.
8-43

-1 50

-1 30

-1 10

-90

-70

-50

.0 01 .1 1 5 10 2 0 35 50 7 0 80 9 0 9 5 9 9

3 MH z

1 MH z

3 00 kHz

1 00 kHz

3 0 kH z

1 0 kH z

10 kHz

30 kH z

1 00 kHz

3 00 kHz

1 M Hz

3 M Hz

r_f1_2 .ezp

Perc ent Exce eding Ordinate

Power (dBm)

Figure 8.84. Electric drill APDs. Figure 8.85. Electric drill detector summary.

-90-80-70-60-50

0 0.005 0.010 0.015 0.020

noise burst every16msec;burst lengthapprox.6msec

driltime.ep

Time(seconds)

) mBd( nroHni a Gi Bd9 no m1t arewoP

Figure 8.83. Typical electrical drill pulse train.

-120-110-100

-90-80-70-60-50-40

1000 2000 3000 4000 500

2

0

0

30405060708090

drilspec.ep

cellular

3 MHz1 MHz300 kHz100 kHz30 kHz10 kHz

Frequency (MHz)

) mBd( m1 @)zHG1tai Bd9. 5( annet nAer utr epA-t natsnoChguor hT,r e woPkaeP

m1ta, m/ VuBd, ht gnert Sdl ei FkaeP

Figure 8.82. Electric drill spectrum as function

of measurement bandwidth.
8-44

8.4 Measurement Conclusions

1. The field strength in a bandwidth derived from an FFT of a full bandwidth pulse shape

measurement matches the field strength shape measured in a narrower bandwidth by a spectrum

analyzer in general shape and absolute amplitude at the peak emission frequency. This confirms

that narrowband spectral measurements with a spectrum analyzer or other measurement receiver

can be expected to provide accurate information to characterize the overall shape and absolute

amplitude of the RF spectrum of UWB signals. 

2. Specialized facilities and expertise are needed to obtain accurate measurements of full

bandwidth pulse shape. Some of the possible errors associated with this technique are subtle, and

many test labs may find that a comparison between FFT-derived spectra and band-limited

spectrum analyzer measurements is the only way to verify the accuracy of the full bandwidth

pulse shape measurements. 

3. There appears to be no simple and accurate technique for relating full bandwidth pulse width

data to the 10-dB or 20-dB emission spectra. This is especially true when the UWB pulse is used

to excite multiple resonant elements, producing an RF pulse with a multi-lobed ringing response. 

4. Amplitude probability distributions (APDs) made with multiple bandwidths provide a useful

and theoretically consistent description of UWB signals and how they would affect receivers

having various bandwidths. Various detector functions (peak, RMS, average voltage, and

average logarithm) can be computed from the APDs or measured directly (if the measurement

instrument has the appropriate capabilities). 

5. For most UWB devices, the recommended FCC Part 15 non-quasi-peak measurement

procedure produces results identical to the average logarithm derived from the 1-MHz bandwidth

APD. In situations where the previous statement is not true, the discrepancy is due to insufficient

video filtering allowing amplitude variations caused by long-duration duty-cycle events (e.g.,

gating) to get through to the peak detector.

6. The value measured with an average logarithm detector is substantially affected by measurement system noise for gated or impulsive signals. The average logarithm value is mostly

insensitive to energy contained in low-duty-cycle, high-amplitude signals. This results in Part 15

measurement values that can be substantially lower (10-15 dB) than the RMS value (average

power) in the UWB signal. 

7. The RMS value produces results that are proportional to measurement bandwidth and spectral

power density, irrespective of PRR or modulation. Depending on the integration time selected

for an RMS measurement, the RMS value can be properly associated with average power when a

gated signal is gated on, average power across an entire gating cycle, or average power during

other operational periods.
8-45

8. Based on measurements of Device C, it seems possible to develop modulation techniques that

could cause a UWB signal to look impulsive/gated in a 1-MHz bandwidth and like a CW signal in

a 50-MHz bandwidth. Such signals could pass the 1-MHz Part 15 average limit (slipping highpower impulsive signals through the Part 15 average logarithm detector) and the 50-MHz 20-dB

peak-to-average limit, while containing average power 15-20 dB above the Part 15 numerical

limits. In fact, a signal could probably be designed so that its average power would be within a

few dB of the peak level permitted in a 50-MHz bandwidth, while still passing the 1-MHz Part 15

average logarithm limit.

9. A Part 15 average limit based on RMS values would not have the problems described in #6,

#7, and #8.

10. Many UWB devices employ dithering, which make their emissions appear more like Gaussian

noise (for bandwidths less than the PRR). While this technique is at least partially successful, the

specific modulation and dithering techniques may leave a spectrum fine structure that contains

many low-level discrete spectral lines and/or other time or frequency distributions that are

significantly different from Gaussian behavior. Some of these non-Gaussian characteristics may

tend to disappear when actual data are transmitted by UWB devices used for communications,

but this effort did not test UWB devices under these conditions.

11. For UWB signals measured with a spectrum analyzer using bandwidths larger than the

average PRR, the peak measured values varied according to a 20 log10 B rule. 

12. Spectrum analyzers having a 50-MHz bandwidth are not yet readily available at test

laboratories. A 20-MHz measurement bandwidth was the largest spectrum analyzer bandwidth

used in these tests to make peak measurements. Based on #11, a useful alternative to making

peak measurements in a 50-MHz bandwidth may be to make measurements in a smaller

bandwidth (but large enough that the UWB signal looks impulsive) and to extrapolate to a 50-

MHz bandwidth using a 20 log10 B rule.

13. We encountered a wide variation of center frequencies (300 MHz to 5 GHz), spectral shapes

(very broad to quite narrow), PRR (10 kHz to 10 MHz), modulation techniques (none, absolute

dither, relative dither, on-off), gating structures (none, 20%-90%), power levels, and operational

functions (short-range to long-range sensing, one-way and two-way communications). 

14. Based on measurements of a small sample of existing UWB devices, we are unable to easily

or confidently assign UWB devices to a limited number of operational categories having certain

distinctive technical characteristics. Therefore, it seems unwise to base any proposed regulations

on the assumption that future UWB devices will limit themselves to a few applications and

techniques that have been currently developed and demonstrated. Instead, it should be assumed

that designers of future UWB devices will creatively explore many additional extensions and

combinations of these technologies, and that any UWB device permitted by the regulations will

eventually be developed -- possibly in large quantities. 
8-46

Figure 8.86. Idealized peak and RMS detector values model.

15. Based on measurements of multiple UWB devices, as well as theoretical considerations, the

following model (Figure 8.86) describes idealized detector response rules for peak and RMS

detectors. This is an idealized treatment, which may or may not apply exactly to specific real

systems. One assumption is that the UWB dithering or modulation produces a signal whose

narrowband statistics are indistinguishable from Gaussian noise. Actual real-world dithering

techniques may or may not approach this ideal, and some UWB devices (e.g., Device C) use

dithering techniques whose narrow bandwidth results are quite non-Gaussian. The model

assumes that the total number of pulses/s remains equal to the PRR; i.e., that neither gating, onoff keying (OOK), or other dithering/modulation techniques significantly changes the total

number of pulses emitted.

Figure 8.86 describes how a UWB signal measured with peak or RMS detectors in a given a

bandwidth can be converted to corresponding readings in another bandwidth. This graph is

referenced to the ratio of receiver bandwidth, B, to pulse repetition rate, PRR. This ratio (rather

than any particular bandwidth or PRR) is what is critical in determining the rules that govern

detector readings. Values of B/PRR >> 1 give non-overlapping impulse responses (the 

impulsive region). Values of B/PRR << 1 give overlapping responses that look like Gaussian
8-47

noise (dithered case) or discrete CW lines (non-dithered case). Values of B/PRR = 1 represent

transitional cases, whose behavior varies depending on the details of the modulation on the PRR. 

The figure shows readings for peak and RMS detectors, for dithered and non-dithered UWB

signals. The figure assumes that no gating is present. If gating is present, the figure should be

modified by reducing all RMS readings by 10 log D, where D is the gating duty cycle, and leaving

all peak readings unchanged. The following notes expand on several of the details of the figure.

Note 1. The dithered RMS detector response strictly follows a 10 log10 B line, since it is assumed

that the total power intercepted by the measurement receiver is strictly proportional to receiver

bandwidth. This will be the case for dithered UWB signals for all B/PRR ratios, as well as for

B/PRR substantially greater than 1 for the non-dithered case, where the number of discrete CW

lines within a bandwidth is proportional to the value of B.

Note 2. The detailed shape of all of these curves near the transition value B/PRR = 1 is not well

known. It will depend substantially on the details of the UWB device modulation and receiver

bandwidth filter response. Filter shapes with sharp suppression of adjacent channel responses

will typically have an extended time response and a smaller peak value (i.e., will appear like a

smaller bandwidth with respect to the peak value and the duration of the pulse). In general, the

slope of the various response curves is trustworthy for bandwidth values far away from the

transition point, but neither the exact shape of the transition curves nor the straight-line intercepts

between the noise-like and impulse-like regions have been precisely determined for a variety of

bandpass characteristics and UWB modulations.. 

Note 3. The behavior near the transition region where B/PRR = 1 for “dithered peak” is particularly obscure. This is partly because there is a transition between a Gaussian probabilistic region

and a deterministic impulsive environment. The area in the shaded triangle represents a range of

responses that has been observed in this set of measurements, but this region should be examined more carefully. Some measured systems show a very tight grouping of detector readings (4-

5 dB difference between average log and peak) when B/PRR = 1, while other systems do not.

Note 4. Since impulses have no time overlap in the impulsive region (B/PRR >> 1), there is no

difference in responses between dithered and non-dithered pulses. Therefore, the dithered and

non-dithered RMS curves will overlay exactly in the B/PRR >>1 region, and both will follow a 10

log10 B slope.

Note 5. For the same reasons described in note 3, the dithered and non-dithered peak curves will

overlay exactly in the B/PRR >>1 region, and both will follow a 20 log10 B slope.

Note 6. The response for non-dithered UWB pulses produces evenly-spaced discrete CW lines

when B/PRR<< 1. The graph assumes that measurements are made centered on one of the

discrete CW lines. Since the envelope response for a single CW line is a constant voltage, the

peak and RMS detectors give the same response. Moreover, since the voltage remains constant 
8-48

for all bandwidths (assuming that the measurement bandpass remains centered on the CW

signal), the common response for peak and RMS detectors follows a single horizontal line for

B/PRR << 1.

Note 7. The value of the dithered peak reading will be about 10 dB above the value of the

dithered RMS reading. This is based on the assumption that dithering produces statistics

identical to Gaussian noise for B/PRR << 1. Also, see Note 8.

Note 8. It should be noted that the ratio between peak and RMS for detected Gaussian noise is

not a constant; instead, it is a statistical quantity that depends on the number of independent

observations. For example, there is a 1% chance that the peak will exceed the RMS by about 7

dB, a 0.1% chance of exceeding the RMS by about 8.5 dB, a 0.01% chance of exceeding the

RMS by about 10 dB, etc. Therefore, the 10-dB peak-to-RMS ratio is based on collecting the

peak reading from 10,000 independent observations. The peak-to-RMS ratio may need to be

adjusted to use other probabilities, depending on the exact circumstances of how the model is to

be applied. 
1The authors are with the Institute for Telecommunication Sciences, National

Telecommunications and Information Administration, U.S. Department of Commerce, Boulder,

CO 80305.

9-1

9. COMPARISON OF THEORY AND MEASUREMENTS

Robert J. Achatz and Roger A. Dalke1

It is important to compare theoretical calculations with measurements whenever possible to

validate results. Such validation is usually based on comparing salient features of measured 

results with corresponding features based on theoretical predictions.

Theoretical results given in Section 3 predict that 1) fixed time-base dithering will attenuate

spectral lines (relative to the non-dithered periodic signal) and also introduce a continuous

spectrum with a shape similar to the pulse spectrum, 2) periodically repeating the dithered signal

will introduce spectral lines at the dither signal repetition frequency, and 3) statistics obtained

from filtered UWB signals will be approximately Gaussian when the bandwidth is less than the

PRR. Note that baseband Gaussian signal statistics yield Rayleigh amplitude statistics.

Device D measurements, summarized in Section 8.3.4, are for a 25% fixed dithered UWB device. 

Measurements include a 10 MHz PRR mode with 100% gating. These measurement results were

compared with theoretical predictions given in Section 3. A synopsis of the comparison is given in

this section. 

Fixed time-base dithering features

Theoretical results for the 50% fixed time-base dithered, 10 MHz PRR PSD are presented in

Section 3.2.1. These results predict that at frequencies of several hundred MHz and higher,

spectral lines are only 10-20 dB above the continuous spectrum and hence should not be

observable when the receiver bandwidth significantly exceeds 20 dB Hz. In addition, the

continuous spectrum should have a shape corresponding to the pulse spectrum. 

The pulse shape of Device D was measured with the full bandwidth test described in Section 5.2. 

The emission spectra was measured with the spectrum analyzer test described in Section 6. The

spectrum analyzer test for a 25% fixed dithered, 10 MHz PRR UWB signal is summarized in

Figure 8.53. These emission spectra measurements have the same shape as the pulse spectrum

measured by the full bandwidth test in Figure 8.49 as predicted. Also, spectral lines at the PRR

intervals were not evident in the emission spectra measurements as predicted by theory.
9-2

Dither signal repetition features

Theoretical results presented in Section 3.2.2 show that for repeated fixed dithered signals,

spectral lines should be observed at frequencies corresponding to the signal repetition period. 

Figure 8.59 shows an emission spectrum measurement of a 25 % fixed dither, 10 MHz PRR

UWB signal with a dither repetition frequency of 10 kHz. The measurement shows the spectral

lines due to the repeating dither signal at 10 kHz intervals as predicted.

Band Limited UWB signal APD features

Theoretical results for the band limited 50% fixed dithered, 10 MHz PRR, UWB signal PSD were

presented in Section 3.3. These results show that when the UWB signal passes through a receiver

filter that is well below the PRR, the signal statistics are approximately Gaussian. As the filter

bandwidth increases, the shape of the density function deviates from Gaussian. This deviation is

measured by a statistic called excess which is a function of filter bandwidth as shown in Figure

3.5. Excesses near zero are indicative of an underlying Gaussian process. The excess factor

deviates from zero at approximately 3 MHz bandwidth. This trend continues and at 10 MHz the

value of the excess is significantly greater than zero.

The APD of for a 25 % fixed dithered, 10 MHz PRR APD shows the same non-Gaussian trend

for bandwidths greater than 3 MHz where the statistics become increasingly impulsive. When the

UWB signal is passed through filters with bandwidths of 1 MHz or less measurements show that

the signal statistics are approximately Gaussian as predicted by theory.