June 2000

NASA/CR-2000-210301

Advanced Atmospheric Water Vapor

DIAL Detection System

Tamer F. Refaat and Hani E. Elsayed-Ali

Old Dominion University, Norfolk, Virginia
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June 2000

NASA/CR-2000-210301

Advanced Atmospheric Water Vapor

DIAL Detection System

Tamer F. Refaat and Hani E. Elsayed-Ali

Old Dominion University, Norfolk, Virginia
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ABSTRACT

ADVANCED ATMOSPHERIC WATER VAPOR

DIAL DETECTION SYSTEM

Measurement of atmospheric water vapor is very important for understanding the

Earth’s climate and water cycle. The remote sensing Differential Absorption Lidar (DIAL)

technique is a powerful method to perform such measurement from aircraft and space.

This thesis describes a new advanced detection system, which incorporates major

improvements regarding sensitivity and size. These improvements include a low noise

advanced avalanche photodiode detector, a custom analog circuit, a 14-bit digitizer, a

microcontroller for on board averaging and finally a fast computer interface.

This thesis describes the design and validation of this new water vapor DIAL

detection system which was integrated onto a small Printed Circuit Board (PCB) with

minimal weight and power consumption. Comparing its measurements to an existing

DIAL system for aerosol and water vapor profiling validated the detection system.
 ii
iii

TABLE OF CONTENTS

Page

LIST OF TABLES.................................................................................................... vi

LIST OF FIGURES.................................................................................................. vii

I. INTRODUCTION................................................................................................ 1

1.1. Atmospheric water vapor.......................................................................... 1

1.2. Water vapor measurement using lidar ....................................................... 2

1.3. Research objectives .................................................................................. 5

II. AVALANCHE PHOTODIODE; THEORY AND MODEL ................................ 8

2.1. Avalanche photodiode structure and theory of operation........................... 8

2.2. Modeling of four layer avalanche photodiode............................................ 11

2.2.1. Rate equations ........................................................................... 13

2.2.2. Continuity equation.................................................................... 16

2.2.3. Avalanche photodiode equivalent circuit .................................... 18

2.3. Model validation....................................................................................... 23

2.3.1. Model parameters ...................................................................... 23

2.3.2. Steady state performance ........................................................... 24

2.3.3. Transient performance................................................................ 31

2.3.4. Model conclusion and limitation................................................. 34

III. AVALANCHE PHOTODIODE; CHARACTERIZATION AND SELECTION. 36

3.1. Experimental setup ................................................................................... 36

3.2. Avalanche photodiode calibration ............................................................. 40

3.3. Temperature dependent responsivity......................................................... 41

3.4. Bias voltage dependent responsivity.......................................................... 45

3.5. Responsivity uniformity scan .................................................................... 49

3.6. Noise measurement .................................................................................. 52

3.7. Frequency response .................................................................................. 54

3.8. Avalanche photodiode selection results..................................................... 56

IV. DETECTION SYSTEM DESIGN AND LABORATORY PERFORMANCE ... 58

4.1. Detection system design .......................................................................... 58

4.1.1. Analog circuit ........................................................................... 61

4.1.2. Digital Circuit ........................................................................... 66

4.2. Detection system settings ......................................................................... 71

4.2.1. Microcontroller program ........................................................... 72

4.2.2. Data acquisition card ................................................................. 74

4.3. Laboratory performance setting ............................................................... 76
iv

Page

4.3.1. Temperature and bias voltage controllers performance ............... 77

4.3.2. Signal conditioning performance ................................................ 79

4.3.3. Digitizer performance................................................................. 81

4.3.4. Overall system performance ....................................................... 83

V. THE LIDAR THEORY ..................................................................................... 87

5.1. The lidar equation .................................................................................... 87

5.1.1. Receiver optics geometry .......................................................... 92

5.1.2. Geometrical form factor for simple overlap ............................... 94

5.2. The DIAL equation ................................................................................. 97

5.3. Applied lidar measurement technique ....................................................... 100

5.3.1. Aerosol measurement ................................................................ 101

5.3.2. Water vapor measurement ......................................................... 102

VI. VALIDATION OF THE DETECTION SYSTEM ............................................ 103

6.1. Receiver system setup .............................................................................. 103

6.2. Detection system validation with NASA airborne DIAL system ............... 106

6.3. Water vapor DIAL detection system validation......................................... 109

VII. CONCLUSIONS ............................................................................................. 113

REFERENCES ........................................................................................................ 116

APPENDICES.......................................................................................................... 125

A. APD MANUFACTURER DATA AND CHARACTERIZATION RESULTS .... 125

A.1. APD designator and manufacturer data ................................................... 125

A.2. SLIK APD package ................................................................................ 126

A.3. APD calibration results ........................................................................... 127

A.4. APD Surface scan results ........................................................................ 128

A.5. APD responsivity variation with temperature .......................................... 129

A.6. APD responsivity variation with bias voltage ........................................... 130

A.7. APD noise measurement ......................................................................... 131

B. DETECTION SYSTEM APPLIED CIRCUITS ................................................. 132

B.1. Analog to digital converter circuit ............................................................ 132

B.2. Data collect and store circuit ................................................................... 132

B.3. Microcontroller connection circuit .......................................................... 133

B.4. Output stage ........................................................................................... 135

B.5. Chip power supply bypassing .................................................................. 136

B.6. Clock and trigger isolation ...................................................................... 136
v

Page

C. ELEMENTS OF THE NEW DETECTION SYSTEM ....................................... 137

C.1. Detection system components ................................................................. 137

C.2. Detection system power consumption ..................................................... 139
vi

LIST OF TABLES

Table Page

2.1 Silicon APD parameters at 300 K. ...................................................................... 23

3.1 APD noise measurements results. ....................................................................... 53

4.1 Timing intervals for different record length......................................................... 76

6.1 Dullas Airport water vapor profile data on November 5, 1999 at 7:00 am........... 11

6.2 Wallops Island water vapor profile data on November 5, 1999 at 7:00 am. ......... 111

7.1 DIAL detection system performance summary.................................................... 115
vii

LIST OF FIGURES

Figure Page

1.1 LASE system block diagram............................................................................... 3

1.2 Schematic of the new DIAL receiver system....................................................... 6

2.1 PIN diode and different APD structures with electric field distribution................ 10

2.2 Schematic diagram of a four-layer APD with electric field profiles...................... 12

2.3 APD equivalent circuit ....................................................................................... 22

2.4 APD spectral response obtained from manufacturer data sheet and model........... 25

2.5 APD quantum efficiency versus wavelength........................................................ 26

2.6 APD responsivity variation with bias voltage at 830 nm...................................... 27

2.7 Responsivity and stability constant variation with multiplication width................ 29

2.8 Donor and acceptor concentration effect on the APD responsivity...................... 29

2.9 Spectral response variation with the width of the absorption region .................... 31

2.10 APD output current with a square wave input power of 100 ps duration........... 32

2.11 APD output current during breakdown............................................................. 33

2.12 APD breakdown time variation......................................................................... 33

3.1 APD experimental setup for the spectral response determination ........................ 37

3.2 APD bias circuit and mounting chamber ............................................................. 38

3.3 APD alignment setup using microscope.............................................................. 39

3.4 Spectral response of the reference detector and test detectors............................. 41

3.5 APD spectral response and responsivity variation with temperature .................... 43

3.6 APD relative error in responsivity due to 1 oC temperature variation................... 45

3.7 APD spectral response and responsivity variation with bias voltage .................... 48
viii

3.8 APD relative error in responsivity due to 1 V deviation in the bias voltage.......... 49

3.9 Responsivity uniformity scan setup..................................................................... 50

3.10 Laser focal point evaluation setup using a pinhole............................................. 51

3.11 APD surface scan............................................................................................. 51

3.12 APD noise-equivalent-power and figure-of-merit measurements ....................... 53

3.13 Frequency response experimental setup ............................................................ 55

3.14 SLIK APD frequency response......................................................................... 57

4.1 Block diagram of the analog circuit .................................................................... 59

4.2 Block diagram of the digital circuit..................................................................... 60

4.3 Circuit diagram of the signal conditioning stage.................................................. 62

4.4 APD voltage bias controller circuit diagram........................................................ 64

4.5 APD temperature controller circuit diagram ....................................................... 65

4.6 Block diagram of the data collect and store circuit.............................................. 67

4.7 System memory map .......................................................................................... 68

4.8 Microcontroller connection block diagram.......................................................... 70

4.9 Block diagram of the output stage...................................................................... 71

4.10 Flowchart of the microcontroller program ........................................................ 73

4.11 Microcontroller ports 1 and 2 configuration...................................................... 73

4.12 Dip switch configuration .................................................................................. 74

4.13 Flowchart of the PCI-DIO-32HS data acquisition card ..................................... 75

4.14 PCI-DIO-32HS data acquisition card connection to the detection system ......... 75

4.15 Detection system data flow diagram ................................................................. 76
ix

4.16 Experimental setup for detection system laboratory performance testing........... 77

4.17 APD temperature controller response............................................................... 78

4.18 APD voltage bias monitor and bias voltage transients ....................................... 79

4.19 Signal conditioning stage normalized gain versus input voltage......................... 80

4.20 Signal conditioning stage frequency response ................................................... 81

4.21 APD and signal conditioning stage low frequency noise spectrum..................... 82

4.22 Histogram plot indicates no missing codes........................................................ 83

4.23 System output versus input light intensity and signal-to-noise ratio ................... 84

4.24 Simulated lidar return signals............................................................................ 86

5.1 Geometry of the target plane at range R ............................................................. 93

5.2 The three possible cases for telescope field of view and laser overlap.................. 96

5.3 Geometry of the biaxial lidar .............................................................................. 96

5.4 DIAL on- and off-line wavelengths relative to water vapor absorption line ......... 98

5.5 Ideal lidar return signal as obtained from the digitizer output .............................. 100

6.1 Water vapor DIAL detection system as a part of the receiver.............................. 104

6.2 Focusing optics alignment setup with the detection system ................................. 105

6.3 Detection system output corresponding to the best alignment ............................. 106

6.4 Detection system validation setup with NASA airborne DIAL system................. 106

6.5 New detection system return signals................................................................... 108

6.6 Aerosol profiles ................................................................................................. 108

6.7 On- and off-line return signals ............................................................................ 110

6.8 Water vapor profiles........................................................................................... 110

6.9 Measured and calculated return signals............................................................... 112
1

CHAPTER I

INTRODUCTION

1.1. Atmospheric water vapor

Water vapor is an important molecular species which is primarily located in the

Earth’s troposphere (part of the atmosphere extending from the surface of the Earth up to

an altitude of about 18 km). Although the distribution of atmospheric water vapor is

highly variable in both time and location, its measurement is very important for

understanding the Earth’s water cycle, the greenhouse effect and weather phenomena.1,2

The water cycle provides a direct way for interactions among Earth’s global systems,

namely, the atmosphere, hydrosphere, cryosphere, lithosphere and biosphere. Water is

considered the main media for energy transfer between most of these systems. Although

the amount of atmospheric water vapor represents only a small percent of the Earth’s

water reservoir, it is very dynamic and its latent heat for transformation is considered the

main energy source that maintains the atmospheric circulation.1,2

Water vapor and clouds affect the incident solar radiation by reflecting solar

radiation back to space and also absorbing some of this energy within the atmosphere.

This substantially moderates the Earth’s climate. On the other hand, water vapor and

clouds affect infrared radiation released by the Earth’s surface. Some of this radiation is

reflected back to the surface and some is absorbed and re-emitted at a lower temperature

which contributes to the global warming problem or greenhouse effect.2

Journal model used for this thesis is Optical Engineering.
2

Water vapor has a direct role in most weather phenomena and natural disasters such

as hurricanes. It was found that the latent heat of water vapor is the main energy source

for hurricanes. The measurement of water vapor flow into a hurricane associated with

other observations aids in estimating the hurricane strength and direction.3,4

Water vapor measurements can be obtained by balloon radio-sondes but a more

effective method is the laser remote sensing technique onboard an aircraft or satellite.

Such systems have the advantage of obtaining two-dimensional water vapor profiles with

high accuracy.5,6

1.2. Water vapor measurements using lidar

Light detection and ranging (lidar) is a remote sensing technique which is currently

used for atmospheric water vapor measurements.5,6 At NASA Langley Research Center,

the lidar atmospheric sensing experiment (LASE) uses the differential absorption lidar

(DIAL) technique to measure water vapor profiles using the ER-2, DC-8 or P3-B

aircraft.6-13 LASE consists of three main systems, the DIAL laser transmitter, the receiver

and the control and data acquisition system.7

In a typical water vapor DIAL system two laser pulses separated by a time interval

are transmitted into the atmosphere. The wavelength of the first pulse is tuned to the peak

of a water vapor absorption line and is called the on-line wavelength, while the second

wavelength is adjusted to a water vapor spectral region with no absorption and is called

the off-line wavelength. The two wavelengths are selected close to each other; therefore,

the scattering by other atmospheric molecules and particles is the same for both on and off

laser wavelengths and the difference in the on- and off-line return signals will be entirely
3

due to the absorption by water vapor molecules. Thus, the ratio of the backscattered

signals at the two wavelengths can be used to calculate the water vapor concentration

profile as a function of range.14-16

This principle was applied to the LASE instrument which is shown in the block

diagram of Figure 1.1. LASE is an instrument that detects water vapor using the 815-nm

water vapor absorption line. LASE measurements were found to have an accuracy better

than 6% or 0.01 g/kg, whichever is greater across the entire troposphere.7-11

Ti:Sapphir

Power Oscillaor

Laser

Diode

Nd:YAG

Pump

ON/OFF

Seeds

TUNABLE LASER

SUBSYSTEM Optics

&

Filters

A

B

C

D

Digirizer

Digirizer

Digirizer

Gain Cont.

APD

APD

88%

12%

HI

MED

LOW

SIGNAL PROCESSOR

Microcontroller

Recoreder

Time Base

Fiber Optic

Laser Return Signal

Output

Tunable Laser Receiver Signal Processor Control & 

Data Acquisition

Figure 1.1 LASE system block diagram.7

The LASE transmitter was designed to generate two laser pulses of 30 ns duration

separated by 400 μs at a frequency of 5 Hz with a pulse energy output of 150 mJ. A

Ti:Sapphire (Ti:Al2O3) power oscillator was constructed using a Nd:YAG laser as the

pump source. Narrow linewidth and wavelength tuning of the Ti:Al2O3 laser is achieved by
4

a continuous wave 100-mW single mode diode laser performing as an injection seed

source. This injected seeding allows control of the spectral linewidth to within 1 pm and

provides wavelength tuning stability to ±0.25 pm. The diode laser wavelength is locked

onto the selected water absorption line and tuned by passing a fraction of its frequency

modulated light through a reference cell filled with water vapor and detecting the cell

transmission. The tunable diode laser seeds the pulsed laser alternately between the online

wavelength located at the center of the water vapor absorption line and the off-line

wavelength typically located 20 to 80 pm away from the online wavelength.7

The LASE receiver has a Cassegrainian telescope with a 40-cm diameter collecting

area. The received light is split into two channels. The two data channels use silicon

avalanche photodiodes (APDs) in order to increase the dynamic range. The first low-gain

channel uses 12 % of the collected light for measuring strong return signals while the

second high gain APD channel uses 88% for low signal level detection. The APD

detectors for both channels where manufactured by EG&G (model C30955E) with a 1.5-

mm diameter active area.7

The APD detector output signal is applied to a trans-impedance amplifier stage

which limits the signal bandwidth to 2.5 MHz. A 1.5 MHz low-pass filter is used to set the

signal bandwidth, and then the signals are applied to 12-bit, 10-MHz digitizers. Both the

amplifier and the digitizer stages are mounted in a CAMAC (Computer Automated

Measurements and Control) crate, which communicates with an onboard computer for

data recording. The whole system is synchronized by a time-based trigger signal.7-9

While LASE has performed well, there are several improvements in the detection

system that could be made to reduce system mass and improve performance. Eliminating
5

the CAMAC crate would reduce the system mass substantially, and using advanced superlow-ionization-coefficient (SLIK) APD technology could reduce the system noise, thus

improving measurement range. These improvements are the subject of this research

dissertation.

1.3. Research objectives

In order to obtain global atmospheric water vapor profile measurements, DIAL

systems such as LASE should be incorporated in aircraft or satellite-based systems. This

requires the development of DIAL systems with reduced size, mass and power

consumption as well as increased measurement capabilities. This will require higher energy

laser systems, but a major part of this development also must be applied to the detection

system.

The research objective of this dissertation was to design, build and test an advanced,

compact, low-mass water vapor DIAL detection system that is used to increase the

performance of the LASE instrument and also serve as a receiver prototype for future

space missions. This new detection system increase the signal-to-noise ratio (SNR) by a

factor of 10 compared to the current LASE receiver. Also it was integrated into a small

size suitable for mounting directly on the receiver telescope. The system consists of an

advanced low noise APD, an analog circuit, a 14-bit digitizer and a digital interface to a

personal computer all on one small electronic card. These goals were achieved using stateof-the-art electronic components and a newly evaluated very low noise APD detector and

constructing a 10-MHz waveform digitizer which is placed as close as possible to this

detector on the same card.17-20 The APD is a very critical device in such systems, since it
6

directly affects the water vapor measurement accuracy and range. Therefore, the selection

of this device was obtained after modeling its performance and characterizing a group of

APDs from different manufacturers.21,22

The design of the new DIAL detection system, the subject of this thesis, is shown

schematically in Figure 1.2. The design allows the placement of both analog and digital

circuits on one small-size light weight printed circuit board which includes detection and

digitization sections. This scheme has the advantage of transmitting digital data for storage

and analysis by a personal computer; therefore, analog signals do not have to be

transmitted for long distances which reduces the system noise pick up.18-20

Optical

Telescope

APD

Detector

Analog

Circuit

Atmospheric

Return

Ligth Signal

Receiver Card

Detected

Signal

Analog

Signal

Optical

Signal

Power Supply Personal Computer System Clock

Digital

Data

Digital

Circuit

Dither Input Laser Fire 

Marker Input Trigger Input

Interface

Figure 1.2 Schematic of the new DIAL receiver system.

The system design was optimized for water vapor absorption lines near 820 and 940

nm. In the receiver system, an existing telescope was used to collect and focus the lidar

return signal onto the APD detector sensitive area. The APD output signal was then

applied to the analog circuit for signal conditioning which includes signal saturation
7

protection, amplification, and filtering. In addition, this stage provides the APD with a

stable bias voltage and temperature control.18-20

The analog signal is then applied to the digital circuit to be digitized and then

processed and temporarily stored. This circuit is compatible with a parallel input/output

computer interface for final data storage and display, and provides monitoring of the

detector high voltage bias and temperature. The whole system card is supplied with ± 5 V

and ± 12 V power and a 10-MHz clock signal for digitization. This new detection system

increases the accuracy and range of water vapor measurements over the current LASE

receiver system.18-20

The new detection system has the advantage of eliminating the CAMAC crate, thus

reducing the mass and power consumption compared to the LASE instrument. A

prototype receiver system was built, tested and evaluated. The prototype receiver system

was used to measure relative aerosol profiles and actual atmospheric water vapor profiles

to validate its performance.23
8

CHAPTER II

AVALANCHE PHOTODIODE:

THEORY AND MODEL

APDs are solid state quantum detectors suitable for low light detection in the visible

and near infrared regions. These devices are commercially available from many

manufacturers and are fabricated using different solid state structures. In this chapter, the

basic APD structures will be described in order to develop a mathematical model for such

devices. The model will be converted into an equivalent circuit and will be validated.

2.1. Avalanche photodiode structure and theory of operation

The basic structure of a PIN diode as well as three different structures of APDs and

their electric field distribution is shown in Figure 2.1. The PIN diode shown in Figure 2.1a

is a p-n junction with an intrinsic or lightly-doped layer sandwiched between the p and n

layers. This structure serves to extend the width of the depletion region which increases

the volume available for absorbing the incident photons. Also it reduces the junction

capacitance, thereby reducing the RC time constant resulting in higher detection

bandwidth. Thus, the response time of these devices is in the tens-of-pico-second range

which corresponds to giga-hertz bandwidths. 24-27 A disadvantage of PIN detectors is that

they have no internal gain mechanism which reduces signal-to-noise ratio. The APD

structure is similar to the PIN diode structure except for an additional gain mechanism

within the device. This gain mechanism depends on the impact ionization process in which

the photoelectric charge carriers, due to their high energy, can impact and cause ionization

of lattice atoms generating more carriers resulting in the internal gain.24-28
9

As shown in Figure 2.1b, the beveled-edge APD has the simplest structure among

them. It consists of a p+-n junction with a high resistivity n layer which increases the

breakdown voltage of the device into the kilovolt range. Breakdown at the edges is

prevented by beveling and making the junction very deep (in the range of 50 μm).

Therefore, the dead part of the p+

 layer is usually etched away to reduce the device depth.

Because the n layer is much deeper than the p+

 layer, electrons produced there are more

likely to be multiplied than holes. This reduces the dark-current noise which is mostly

generated by hole current. The disadvantage, however, is that only light absorbed in the p

layer leads to effective multiplication and this layer has the lowest electric field. As a

result, charge accumulates slowly leading to a longer response time typically in the range

of tens of nanoseconds.28

The reach-through structure APD and its electric field distribution are shown in

Figure 2.1c. This structure consists of an absorption region and a separate multiplication

region. In the absorption region, the p+

 layer at the active surface is followed by an

intrinsic wide layer which increases the photon absorption depth. The emitted photoelectrons drift and reach a constant velocity. In the multiplication region, the p-n+

 layers

form a thin junction with a high internal field which enhances the impact ionization

process.28-30 Recently, an enhancement of the reach-through structure known as the “super

low ionization ratio κ” (SLIK) geometry has become available and is shown in Figure

2.1d.31 The ability of electrons and holes to “impact ionize” in order to generate additional

charge carriers is characterized by the ionization coefficients α and β, for
10

electrons and holes respectively. The ionization coefficient represents the ionization

probabilities per unit length. An important parameter is the ionization ratio, κ, given by

κ β

α = . (2.1)

(a) P-I-N diode structure and 

 electric field distribution.

(b) The beveled edge or deep 

diffusion APD structure and 

electric field distribution.

(c) Reach through APD structure and 

 electric field distribution.

(d) SLIK APD structure and 

 electric field distribution.

Electric Field

p n i

Electric Field

p n

Electric Field

p+ i p n+

Electric Field

p+ i p n+

+

Figure 2.1 PIN diode and different APD structures with electric field distributions.

In APDs, it is desirable to reduce κ as much as possible so the multiplication process is

only due to electrons, since hole current leads to high dark-current noise. 24 This was

achieved in the SLIK structure by increasing the absorption region electric field and
11

merging this region with the multiplication region.31 Typically, the value of κ is in the

range of 0.02 for the reach-through structure while it is 0.002 for the SLIK structure.31

2.2. Modeling of four layer avalanche photodiode

Large efforts were made to model the different structures of APDs, but little work

has focused on developing an equivalent circuit for these devices.28,32 Chen and Liu

constructed an equivalent circuit model for three layer PIN-APD and applied it to

In0.53Ga0.47As-InP PIN-APD.33 This model was based on the study of the minority carrier

generation, diffusion and drift through the different layers of the device, by applying both

the rate and continuity equations. The drift-diffusion model was converted into an

equivalent circuit. In this section the, Chen-Liu model will be extended to the case of four

layer APDs, such as the reach-through and SLIK structures, in order to solve for the

characteristics and response and to obtain an equivalent circuit for such devices.21 This

APD model is capable of simulating the transient and steady state behavior of an APD

knowing its microstructure. The model simulation results were compared to a commercial

reach-through silicon APD in order to validate it.21,34

A schematic of the structure of a reach-through APD as well as the internal electric

field distribution are shown in Figures 2.2a and 2.2b respectively. As discussed earlier, the

field profile shown in Figure 2.2b distinguishes the two main regions within the device.

The first is the absorption region which is used to absorb the incident photons to generate

photo-charge carriers and drift these carriers into the second region. The second region is

the multiplication region which has a relatively smaller depth with higher field. In this

region, the photo-generated carriers produce additional carriers by the impact ionization
12

process. For the sake of simplicity in this model, the electric field profile will be

approximated by two step functions as shown in Figure 2.2c. The lower field value is for

the absorption region that is covered by the i layer, while the higher field value is for the

multiplication region which is almost covered by the p layer. The electric field will be

neglected in the outer highly-doped layers since they have relatively smaller depletion

widths. Under these approximations the device currents in the p+-i and p-n+ interfaces will

be due to the diffusion currents of the minority carriers form the highly doped layer, and

the drift current of the generated charge carries in the internal regions.21

G1

Wpp

p i p + n+

Go

Gpp

Gi

Gp

Gnn

Wi Wp Wnn

Lpp Lnn

Ei

Ep

E-Field

Depth

(a)

(b)

(c)

E-Field

Depth

Figure 2.2 (a) Schematic diagram of a four layer APD showing light absorption through

the device with (b) simulated34 and (c) approximate electric field profiles.
13

The absorption profile along the device depth is also shown in Figure 2.2a.

Considering the input optical power Pin at a wavelength λ, the incident photon flux is

given by

= λ

hc

P G in

o , (2.2)

where h is Plank’s constant and c is the speed of light in vacuum. A part of this flux will

reflect back at the APD surface and the other part will penetrate the device and will be

given by

G1 = Go ( ) 1− R , (2.3)

where R is the APD surface reflection coefficient. Since the electron-hole pair photogeneration rate is directly proportional to the number of absorbed photons, this rate will be

given as (in s-1) for the p+, i, p and n+ layers, respectively, by

[ ( )] pp 1 ppWpp G = G ⋅ 1− exp − α (2.4)

( ) [ ] ( ) i 1 pp pp iWi G = G ⋅ exp −α W ⋅ 1− exp − α (2.5)

( ) [ ( )] p 1 pp pp i i pWp G = G ⋅ exp −α W − α W ⋅ 1− exp − α (2.6)

( ) [ ] ( ) nn 1 pp pp i i p p nnWnn G = G ⋅exp − α W − α W −α W ⋅ 1− exp −α , (2.7)

where αpp, αi, αp and αnn are the absorption coefficients and Wpp, Wi, Wp and Wnn are the

widths of the p+, i, p and n+ layers, respectively.

2.2.1. Rate equations

Considering a reverse bias device, the rate of increase of the number of electrons in

the p+ layer (minority carriers) will be proportional to the photo-generation rate and
14

opposed by the recombination and diffusion. Thus the rate equations for the electrons in

the p+ layer is given by

q

P I

G

dt

dP ppe

ppe

ppe

pp

ppe − τ = − . (2.8)

Similarly for the n+ layer, the rate equation of the holes (minority carriers) is given by

q

P I G

dt

dP nnh

nnh

nnh

nn

nnh − τ = − , (2.9)

where Pppe and Pnnh are the number of excess electrons and holes, τppe and τnnh are the

electron and hole life times for the p+

 and n+ respectively, Ippe is the electron diffusion

current from the p+

 to the i region, Innh is the hole diffusion current from the n+ to the p

region and q is the magnitude of the electron charge.

In the i region, both electrons and holes are considered as minority carriers, and the

rate of increasing any type of these carriers will be equal to the photo-generation rate in

this region plus the impact ionization rates initiated by any electrons or holes minus the

recombination rate. Also, for electrons this process will be affected by the electron

diffusion and drift from the p+ region and to the p region respectively. For holes, it will be

affected by the hole drift from and to the p and p+ regions respectively. Therefore, the rate

equation for electrons in the i region has the form

iet

ie

ier

ppe ie

i ie ie ie ih ih ih

ie P P

q

I

G v P v P

dt

dP

τ − τ = + ζ + ζ + − , (2.10)

where vie and vih are the electron and hole velocity in the i region, ζie and ζih are the

electron and hole impact ionization rates and τier and τiet are the electron recombination
15

life-time and electron transient time in the i region respectively. For holes, the rate

equation is

iht

ih

ihr

ih

pht

ph

i ih ih ih ie ie ie

ih P P P G v P v P

dt

dP

τ − τ − τ = + ζ + ζ + , (2.11)

where τpht, τihr and τiht are the hole transient time in the p region, hole recombination life

time in the i region and the hole transient time in the i region respectively. It should be

noted that equations (10) and (11) are equal due to the charge neutrality condition. For

the reach-through APD, the impact ionization terms can be neglected and the photogeneration and drift terms are the dominant. This assumption is not valid for the SLIK

structure APD since the electric field in that region is relatively higher causing the impact

ionization to occur earlier.31

Finally, since the p layer is lightly doped, both electrons and holes will be considered

as minority carriers, and the rate equations will be similar to that of the i region and are

given by

pet

pe

per

pe

iet

ie

p pe pe pe ph ph ph

pe P P P

G v P v P

dt

dP

τ − τ − τ = + ζ + ζ + (2.12)

and

pht

ph

phr

nnh ph

p ph ph ph pe pe pe

ph P P

q

I G v P v P

dt

dP

τ − τ = + ζ + ζ + − , (2.13)

where ζpe and ζph are the electron and hole impact ionization rates in the p region, τiet, τper

and τpet are the electron transient time in the i region, recombination life-time in the p

region and transient time in the p region respectively and τphr is the hole recombination

life-time in the p region. Also, due to the charge neutrality condition, equations (2.12) and
16

(2.13) are equal. The dominant terms in these relations will be the impact ionization and

the drift terms for both the reach-through and SLIK APDs, while the absorption term can

be neglected especially for deep devices.

2.2.2. Continuity equations

The diffusion current from the p+

 layer can be obtained by applying the continuity

equation which is given by

( ) ( ) 0

A

n n g x

dx

d n D pp

ppe

e eo

2

e

2

e + = τ

− − , (2.14)

where De is the electron diffusion coefficient in the p+

 region, ne is the electron density

profile, neo is the electron equilibrium density and A is the device area. The boundary

conditions for this layer are; at x = 0, ne = ne(0) and at x = Wpp, ne = 0. If Lpp is the

diffusion depth in the p+

 region, the diffusion coefficient is given by

ppe

2

pp

e

L

D

τ = , (2.15)

and gpp(x) is photo-generation rate of the charge carriers along the depth of the p+ layer.

Referring to equation (2.3), gpp(x) is given by

g ( ) x G exp( x) pp 1 pp − αpp = ⋅α ⋅ 0 x Wpp ; ≤ ≤ . (2.16)

The solution of equation (2.14) is given by

( ) ( ) [ ] [ ] ( )

































− − −α































 −

= + −α + − −

pp

pp

pp

eo 1 pp pp

pp

pp

pp

pp

e eo 1 pp e eo 1

L

W

sinh

L

x sinh

n c exp W

L

W

sinh

L

W x

sinh

n x n c exp x n (0) n c

 (2.17)

where c1 is a constant given by
17

( )

( ) 2 in

pp

2

pp

pp ppe

1 P

Ahc 1 L

1 R

c −α

− λα τ = . (2.18)

To evaluate ne(0), equation (2.17) is applied to the condition

( ) ∫ = −

Wpp

0

ppe e eo P , (2.19) A n n dx

and solving for ne(0) and substituting its value in equation (2.17), the electron diffusion

current from the p+ region can be determined from

Wpp x

e

ppe e dx

dn I qAD

=

= + . (2.20)

Similarly, to evaluate the hole diffusion current Innh from the n+ region, we apply the

hole continuity equation which is given by

( ) ( ) 0

A

n n g x

dx

d n D nn

nnh

h ho

2

h

2

h + = τ

− − , (2.21)

where Dh is the hole diffusion coefficient in the n+ region, nh is the hole density profile and

neo is the hole equilibrium density. The boundary conditions in this region are at x = 0, nh =

0 and at x = Wnn, nh = nh(0). The hole diffusion coefficient can be obtained using the

diffusion length Lnn in the n+

 region by

nnh

2

nn

h

L D

τ = , (2.22)

and gnn(x) is photo-generation rate of the charge carriers along the n+ layer depth.

Referring to (2.3) gnn(x) is given by

gnn ( ) x = G1 ⋅exp(− αppWpp − αiWi − αpWp )⋅αnn ⋅ exp( ) − αnn x Wnn ; 0 ≤ x ≤ (2.23)

The solution of equation (2.21) is:
18

() ( ) ( ) [ ]

































= + − α + − − − α

nn

nn

nn

h ho 2 nn h ho 2 nn nn

L

W sinh

L

x sinh

n x n c exp x n (0) n c exp W

[ ]































 −

− −

nn

nn

nn

nn

ho 2

L

W sinh

L

W x sinh

n , (2.24) c

where c2 is a constant given by

( )

( ) ( ) 2 pp pp i i p p in

nn

2

nn

nn nnh

2 exp W W W P

Ahc 1 L

1 R

c −α −α −α −α

− λα τ = . (2.25)

Similarly, to evaluate nh(0) we use

( ) ∫ = −

Wnn

0

Pnnh A nh nho dx . (2.26)

Substituting its value in equation (2.21), we get a complete expression for the hole density

in the n+. The hole diffusion current will then be evaluated using

x 0

h

nnh h dx

dn I qAD

=

= − . (2.27)

2.2.3. Avalanche photodiode equivalent circuit

For the purpose of constructing an equivalent circuit for the APD, we assume a

constant Co which is equivalent to a capacitor. Referring to equations (2.4), (2.5), (2.6)

and (2.7), equations (2.8), (2.9), (2.10) and (2.13) can be written respectively as

ppe

pp

pp pp

o

opp

in

opp I

R

V

dt

dV

C

V

P I = = + + (2.28)
19

nnh

nn

nn nn

o

onn

in

onn I

R

V

dt

dV C

V

P I = = + + (2.29)

ppe is

it

i

ir

i i

o

oi

in

oi I I

R

V

R

V

dt

dV C

V

P I = = + + − − (2.30)

nnh ps

pt

p

pr

p p

o

op

in

op I I

R

V

R

V

dt

dV

C

V

P I = = + + − − , (2.31)

where the excess charge carriers are converted to voltages, using Co, and are given by

o

ppe

pp C

P

V = q , 

o

ie

i C

P V = q , 

o

ph

p C

P

V = q , and 

o

nnh

nn C

P V = q (2.32)

The rest of the parameters are given by

pp

in

opp G

P

q

1 V = ⋅ , 

i

in

oi G

P

q

1 V = ⋅ , 

p

in

op G

P

q

1 V = ⋅ and 

nn

in

onn G

P

q

1 V = ⋅ (2.33)

o

ppe

pp C

R

τ = (2.34)

o

nnh

nn C

R τ = (2.35)

o

ier

ir C

R τ = ,

o

iet

it C

R τ = (2.36)

( ) is o i ie ie ih ih I = C V v ζ + v ζ (2.37)

o

phr

pr C

R

τ = ,

o

pht

pt C

R

τ = (2.38)

( ) ps CoVp vpe pe vph ph I = ζ + ζ . (2.39)

The diffusion currents Ippe and Innh will be given respectively by

p in ppeo

pd

pp

ppe P I

R

V

I = + β + (2.40)
20

n in nnho

nd

nn

nnh P I

R

V I = + β + , (2.41)

where















 −













 = 1

L

W

R R cosh

pp

pp

pd pp (2.42)





 



 −













 = 1

L

W R R cosh

nn

nn

nd nn (2.43)

( )

( ) ( ) [ ] ( )































 −















α

− α − α − α + − αα

λ − β =

1

L

W

L cosh

exp W 1

exp W

1 L

L

hc

1 R

q

pp

2 pp

pp pp

pp pp

2 pp pp pp

pp

2

pp

2

pp pp

p

( )

































− α

















+















+

pp

pp

pp

pp pp

pp

pp

L

W

L sinh

1 exp W

L

W

cosh

 (2.44)

( )

( ) ( )



































 





+















− α − α − α − α

α λ − β =

nn

nn

nn

nn

nn

2 pp pp i i p

nn

2

nn

2

nn nn

n

L

W L sinh

1

L

W cosh

exp W W Wp 1 L

L

hc

1 R

q

[ ] ( )















− α





 



 −













 α

− α − + nn

nn

2 nn

nn nn

nn nn

1

L

W L cosh

exp W 1 (2.45)
21



























































 +

=

pp

pp

pp

pp

pp

e eo

ppeo

L

W

sinh

L

W

1 cosh

L

AD n I q (2.46)



























































 +

=

nn

nn

nn

nn

nn

h ho

nnho

L

W sinh

L

W 1 cosh

L

AD n I . (2.47) q

Using equation (2.28) through (2.31) and the definition of the diffusion currents

given in equations (2.40) and (2.41), the APD equivalent circuit can be constructed as

shown in Figure 2.3. The APD output current, IAPD, will be given by

( ) d

BIAS

APD ppe i p s j I

dt

dV I = I + I + I + C + C + , (2.48)

where

it

i

i R

V I = (2.49)

pt

p

p R

V

I = , (2.50)

and Cs and Cj are the parasitic and junction capacitance, Id is the sum of the leakage and

tunneling currents and VBIAS is the applied APD bias voltage.
22

Iopp

Ionn

Ioi

Iop

P in

P in

P in

P in

V opp

V onn

V op

V oi

Iopp

Ionn

Ioi

Iop

C o

C o

C o

C o

R pp R pd

R nn R nd

Ippe

Innh

Ippeo

Innho

Innh

Ippe Iis

Ips

pP in B

R ir R it

R pr R pt

Ii

Ip

V i

V p

V pp

V nn

C s C j V BIAS

IAPD

Id Ippe Ii Ip

nP in B

Figure 2.3 APD equivalent circuit
23

2.3. Model validation

In order to validate the discussed model, a silicon reach-through APD was

simulated. The APD is manufactured by EG&G (model number C30902E). Table 2.1

gives some parameters of this APD as well as for silicon at a temperature of 300 K.30,35-40

Table 2.1 Silicon APD parameters at 300 K.

Parameter Value

A(mm2) 0.196

Wpp(μm) 1

WI(μm) 15

Wp(μm) 4.5

Wnn(μm) 10

R 0.35

NA(cm-3) 5x1012

ND(cm-3) 5x1012

NI(cm-3) 1.5x1010

De(cm2s

-1) 34.2

Dh(cm2s

-1) 11.9

τppe(μs) 30

τnnh(μs) 20

τier(μs) 30

τphr(μs) 20

2.3.1. Model parameters

The values of the electric field were approximated by two step functions under

constrain of equal bias voltage for both the simulated and approximated profiles shown

earlier in Figure 2.2b and 2.2c. The electric field step function relates to the applied bias

voltage by

( ) i BIAS i 5Wp E = V W + 6. and p 5Ei E = 6. , (2.51)
24

where Ei and Ep are the electric in the i and p layers respectively. Since the device built-in

potential is neglected, these relations are only valid near the breakdown bias voltage of the

APD.

The impact ionization coefficients for electrons and holes as a function of the electric

field in the range of 2x105

 to 5x105 V/cm were obtained form ref. 41 and applied for the i

and p regions.

The electron and hole drift velocity can be obtained as a function of the electric field

from the empirical formula37

[ ] ( ) β β +

= 1

C

C

S

1 E E

E E v , (2.52) v

where vS is the carrier saturation velocity, EC is the critical field defined by the ratio of the

carrier saturation velocity to its mobility and β is a constant. The values of the saturation

velocity, mobility and the constant β are 1x107

 and 0.8 x107 cm/s, 1320 and 460 cm2/Vs

and 0.9 and 0.8 respectively for electrons and holes.

The absorption coefficients for the various layers where assumed the same. The

absorption coefficient as a function of the wavelength of the incident light was taken from

ref. 42 for the 800 to 1100 nm wavelength range. The data where fitted using polynomial

fit and applied to the model.

2.3.2. Steady state performance

The spectral response of the APD under investigation was obtained form the

manufacturer data sheet and is shown in Figure 2.4 at a temperature of 295 K.34 Also

shown in the same figure the spectral response obtained after applying the model to the
25

same device assuming a 215 V bias voltage and 300 K temperature. Comparing the two

curves we note that the model spectral response has a lower responsivity values with a

maximum near the 870 nm. The responsivity R was obtained using the relation.

in

APD

P

I R = . (2.53)

800 850 900 950 1000 1050

10

20

30

40

50

60

70

80

Wavelength ( nm )

Responsivity ( A/W )

Manufacturer Data Sheet

Model Result

Figure 2.4 APD spectral response obtained from the manufacturer data sheet

at 295 K and from the model at 300 K and 215 V.

The quantum efficiency variation with wavelength is shown in Figure 2.5 obtained

from the manufacturer and the model.21,34 In order to obtain this relation, the model was

run while setting the impact ionization coefficients to zero. Therefore, the responsivity

gave a measure to the quantum efficiency η according to the relation.

G 1 q

hc R = λ η = (2.54)
26

800 850 900 950 1000 1050

0

10

20

30

40

50

60

70

80

90

100

Wavelength ( nm )

Quantum Efficiency

Manufacturer Data Sheet

Model Result

Figure 2.5 APD quantum efficiency versus wavelength at obtained form the manufacturer

and form the model at 300 K and 215 V bias.

Although the results of figures 2.4 and 2.5 indicate the lack of accuracy in the model

parameters, the device gain variation with respect to wavelength can be predicted at the

given operating voltage. The accuracy of the model parameters is very critical in the model

stability and results, especially in the multiplication region, as will be shown later.

Figure 2.6 shows the responsivity variation with bias voltage obtained form ref. 34

and the model both assuming constant wavelength of 830 nm.21,34 The model result

indicates that the APD bias voltage of 215 V is near-breakdown voltage which is about 8

V less than the manufacturer corresponding voltage at similar temperature.

As shown in Figure 2.6, the large deviation between the simulated results and the

manufacturer data at low operating bias voltage was expected. This was indicated earlier

in equation (2.51) since the natural built-up potential is neglected. This affects the impact

ionization rate leading to decrease of the responsivity at lower voltage biases.
27

150 160 170 180 190 200 210 220

0

20

40

60

80

100

120

Bias voltage ( V )

Responsivity ( A/W )

Manufacturer Data Sheet

Model Result

Figure 2.6 APD responsivity variation with bias voltage at 830 nm obtained from the

manufacturer data sheet at 293 K and from the model at 300 K.

To demonstrate the model sensitivity to the device multiplication width, Wp,

referring to (2.13) we can define a multiplication stability constant CMS as

phr pht

MS ph ph pe pe

1 1 C v v

τ − τ = ζ + ζ − , (2.55)

in which the electron impact ionization and the hole transient time terms are the dominant.

The value of this constant determines the stability of the device since it controls the

feedback of the multiplication process. Figure 2.7 investigates the variation of the stability

constant and the resulted responsivity with the width of the multiplication region Wp. The

values of the multiplication stability constant shown to the left of the stability limit line are

positive while the values to the right are negative. Wp affect the hole transient time

according to

ph

p

pht v

W

τ = . (2.56)
28

It was found that any positive values of CMS causes model failure because it causes an

unstable positive feedback. Although the magnitude of the negative values of CMS controls

the APD gain and thus its responsivity, the practical value of the responsivity lies within a

small region of the multiplication width. The bias voltage is another factor affecting the

multiplication stability constant, since it controls both the charge carrier velocity and

impact ionization coefficients. This explains the sudden increase of the responsivity with

bias voltage as shown in Figure 2.6.

Practically, when manufacturing these devices, it is very hard to keep the

multiplication width consistent. This explains why similar APD part numbers from the

same manufacturer have different operating bias voltages with different characteristics.22

The value of the responsivity is also affected by the initial doping concentration of

the donor atoms in the n+

 region. As shown in Figure 2.8, at a constant acceptor

concentration, increasing the doping level of the donors reduces the responsivity. A

doping level of 5x1012 is a reasonable limit for manufacturing devices with similar

characteristics as the device under investigation. Shown in the same figure the effect of the

acceptor atoms doping concentration in the p+

 region, at constant donor concentration,

which has a smaller effect.
29

4.4 4.5 4.6 4.7 4.8 4.9 5

105

106

107

108

101

102

103

104

|C | ( 1/s ) MS

Responsivity ( A/W )

Multiplication Region Width ( μm )

R=∞

Stability Limit

Multiplication Stability Constant

APD Responsivity

Figure 2.7 Responsivity and multiplication stability constant variation with the

multiplication width, obtained at 800 nm wavelength and 215 V bias.

1010 1012 1014 1016

60

80

100

120

140

Doping Concentration (1/cm )

Responsivity ( A/W )

3

Donor Level Variation

Acceptor Level Variation

Figure 2.8 Donor and acceptor concentration effect on the APD

responsivity at 800 nm wavelength and 215 V bias voltage.

As pointed out by applying the continuity equation, increasing the donor

concentration increases the electron diffusion current to the absorption region. This has
30

the effect of increasing the recombination rate with holes leading to decrease the photo

generated carries (holes) resulting in responsivity reduction. Increasing the donor level

more, responsivity saturation will be resulted, since the diffused electrons recombined with

all the generated holes. A Similar effect occurs due the increase of the acceptor level.

However, since the acceptor diffusion current take place in the multiplication region where

the photo charge carriers got multiplied, it has a lower effect on the responsivity.

One application of this model is to study the effect of different APD parameters on

its characteristics. This is demonstrated in Figure 2.9, which gives the effect of changing

the width of the i or absorption region on the spectral response. By increasing the

absorption depth the spectral response curve shifts to the infrared. Referring to equation

(2.51), the degradation of the peak points is due to the considerable reduction in the

electric field of the absorption region, Ei, since the bias voltage was assumed constant

(215 V) for all cases. This spectral response shift is known commercially as infrared

enhanced APDs.

The value of the absorption width, Wi, is limited by two main factors. Reducing Wi

to a low value limits the absorption depth and the acceleration distance required by the

charge carriers to gain sufficient energy for the impact ionization process. On the other

hand, increasing W to large value causes losses in the carriers due to recombination.
31

800 850 900 950 1000 1050

0

10

20

30

40

50

60

70

Wavelength in nm

Responsivity in A/W

W = 15 um i

W = 35 um i

W = 55 um i

W = 75 um i

W = 100 um i

Figure 2.9 Spectral response variation with the width

of the absorption region obtained at 300 K and 215 V.

2.3.3. Transient performance

To test the APD model in the transient state, an input optical pulse was simulated

and applied to the model. This pulse had a square waveform with 100 ps duration and 1

μW amplitude at 800 nm. The APD output current is shown in Figure 2.10 for different

operating bias voltage. The rise time was observed to be constant at about 80 ps. The fall

times were 450 and 780 ps for 140 and 215 V bias voltages respectively. This indicates

that the rise time has a negligible variation with bias voltage, while the fall time increases

with the increase in the voltage. The rise time generally depends on the speed of the

photon absorption process within the device; therefore, it has nothing to do with the bias

voltage. On the other hand, the bias voltage affects the speed of the charge carriers and

their number. That’s why it affects the fall time. The sudden changes in the APD current
32

shown in Figure 2.10 at the rising and falling instances are due to the sharp edges of the

square wave of the simulated input power.

0 0.2 0.4 0.6 0.8 1

0

50

100

150

200

250

 Time ( ns )

 APD Current ( nA )

V = 215 V BIAS

V = 140 V BIAS

Figure 2.10 APD output current with a square wave input power of 100 ps duration

and 1 μW amplitude at 800 nm, obtained at 140 and 215 voltage biases.

This model is also suitable for performing some tests to the APD which are

practically destructive to the device. Figure 2.10 shows the simulated APD output current

during breakdown. This was simulated assuming an applied bias voltage of 215.7 V and a

step input power of 1 μW as shown in the same figure. The breakdown time, shown in the

same figure, was defined as the time interval between applying the input power and the

rapid increase of the APD current up to 10% of its normal value at starting. The 10 %

value was suitable enough since the APD current has a rapid increase to infinity indicating

device damage. Figure 2.11 shows the relation between the magnitude of the breakdown
33

voltage and the breakdown time indicating that increasing the applied breakdown voltage

speeds up the breakdown time. APD Current ( 

μA )

35

40

45

50

55

0 20 40 60 80 100

0

1

2

Input Power ( 

μW )

Time ( μs )

Breakdown Time

10%

Figure 2.11 APD output current during breakdown simulated with an applied

voltage bias of 215.7 V and a step input power of 1 μW. The breakdown time is

defined as the time interval between applying the input power and the increase

of the APD current up to 10% of its steady-state value.

215.4 215.6 215.8 216 216.2 216.4 216.6

20

40

60

80

100

120

Breakdown Voltage ( V )

Breakdown time ( 

μs )

Figure 2.12 APD breakdown time variation with breakdown voltage bias.
34

For low light detection, an APD can be operated in the Geiger mode or photon

counting.31,43-46 In this mode, the device is biased with a voltage above its breakdown

value leading to increase the device gain. Therefore, it will be very sensitive and any

absorption of a small amount of photons will cause avalanche breakdown which can be

detected as current pulse. In this mode of operation, it is important to apply such overbreakdown bias for short time intervals known as the counting bin to control the avalanche

breakdown. Increasing the counting bin will cause damage to the device.31

Although the switching action was assumed to the input power, the characteristics

shown in Figure 2.11 is important for defining the bin duration for an APD operating in

the Geiger mode. With the current model, switching the bias voltage was difficult to

obtain. This is due to the dependence of many parameters on the bias voltage which had to

be calculated before starting the simulation.

2.3.4. Model conclusion and limitations

A drift-diffusion model for four layer APDs has been developed. This model is a

modification of the Chen and Liu PIN-APD model. The model was used to drive an APD

equivalent circuit and can be used to simulate the device transient and steady state

response. For the validation purpose the model parameters were chosen as close as

possible to an actual device. Reasonable agreement between the model results and the

manufacturer data sheet indicated the validity of the model. This model can be used for

device structure modification as to enhance the spectral response of a device or for

destructive tests such as breakdown.21
35

There were many assumptions associated the construction of this model which make

it limited. The limitations of the model include the effect of temperature, low bias voltage

operation, simulation of the device characteristics and response near saturation and noise

modeling.
36

CHAPTER III

AVALANCHE PHOTODIODE:

CHARACTERIZATION AND SELECTION

A group of silicon based avalanche photodiode (APD) detectors, with different

structures, were characterized from different manufacturers. The main concern in these

experiments was to calibrate these APDs and to investigate the influence of their high

voltage bias and temperature on the APD responsivity and to examine their active area

uniformity and noise performance.22 With these results, an optimal detector was chosen

for the new atmospheric water vapor DIAL detection system. The detection system can

operate at 720, 820 or 940 nm wavelengths which correspond to water vapor absorption

lines.18,22,47

The APDs that were characterized are given in Appendix A as well as a summary of

the characterization results. The results of the reach-through structure (RTS) APD

C30955E48 currently used in the LASE instrument and the super low ionization coefficient

(SLIK) APD C30649E49 will be discussed in detail in this chapter. Both APDs are

manufactured by EG&G.47-49

3.1. Experimental setup

The experimental setup for the APD responsivity calibration is shown in Figure 3.1. The

light source was a broad-band halogen lamp supplied by a stabilized power supply to

insure stable spectrum and intensity. The lamp output is filtered by a 600-nm long-pass

filter to prevent higher order dispersion of shorter wavelengths from being collected in
37

first-order dispersion in the range of 600 to 1100 nm. The chopper was used to modulate

the optical signal for dc offset elimination at the detector output and to allow for phaselocked detection. The monochromator was used to separate the light input into its spectral

components. An integrating sphere was used to diffuse the exiting light to insure uniform

intensity at the detector.22,45,47,50

Light

Source

600 nm

Monochrometer Optical Filter

Integrating

Sphere

Detector

Chamber

Output Signal

Optical

Chopper

Test APD

Oscilloscope

Lockin Amplifier

Voltage Bias

& Power Supplies

Temperature

Controller Digital Voltmeter

RC High

Pass Filter

Computer

Control

Chopper

Controller

Temperature Signal

Synchronizing Signal

Monochrometer Control

Figure 3.1 APD experimental setup for the spectral response determination.

The APD output was filtered by a high-pass filter to eliminate dark current and

background radiation offsets. An oscilloscope was used to check the detected signal and

to obtain its peak-to-peak value, Vp-p, and a lock-in amplifier was used to accurately

measure the signal rms value, Vlock-in. The chopper controller adjusted the chopping

frequency to 200 Hz and supplied synchronization signals for the other instruments. 22,45

The APDs are biased using a high voltage source. The APD current variation,

representing the change in the light intensity, was converted to a voltage signal using a
38

trans-impedance amplifier (TIA).24-27,51-52 The TIA configuration is shown in Figure 3.2a,

where Rf is the amplifier feedback resistance and R and C act as a low pass filter to

eliminate any bias voltage ripples. The test detector was placed on a printed circuit board

and put inside a chamber, as shown in Figure 3.2b. The chamber was located on a threedimensional, computer-controlled, translation stage for alignment. Nitrogen gas flow was

used to prevent condensation of water vapor on the detector window at low temperature

and to avoid dust accumulation. A temperature controller and a thermoelectric cooler

(TEC) were used to fix the temperature of the APD under test.53 Water circulation

provided forced cooling to the TEC for low temperature settings. The APD temperature

was measured with a temperature sensor and a digital voltmeter. A personal computer sent

commands to the monochromater to adjust the wavelength for the spectral scan, and it

acquired the lock-in amplifier and the temperature readings using a GPIB data acquisition

card.22,47

(a) (b)

Water

Circulation N Flow 2

TEC

Window

APD

Heat

Sink

Detector Chamber

Electrical

Connections

High Voltage

+

-

3

2

6

R R

C

f

APD

Light Input

Output

Signal

TIA

High

Voltage

Figure 3.2 (a) APD bias circuit and the trans-impedance amplifier connection

(b) APD chamber used to mount and stabilize the device temperatures.
39

The APD spectral response was measured over a 600 to 1100 nm wavelength range.

Comparison was made with a NIST (National Institute of Standards and Technology)

calibrated reference detector by placing each detector in the same uniform light field at the

same position. The distance between the light outlet and the detector active area was 150

mm. For alignment purposes, a microscope with a 200 μm depth of focus was used to

position all detectors as shown in Figure 3.3. Applying the inverted square function, the

worst-case deviation of the intensity at the detector was ±0.53 %. Errors due to

positioning of the detectors can cause absolute calibration uncertainty of less than 1%.

The microscope was placed on a kinematic mount (1μm placement precision) so that it

could be removed from the optical path during the wavelength scan and precisely replaced

in the path for detector positioning.22,47

Light

Source Reflector

Eye Piece

5x Objective

lens

Auxiliary

Light Source

Optical Axis

Detector

x

y

z x

Intensity Adjustment

Microscope

Figure 3.3 APD alignment setup using microscope.
40

3.2. Avalanche photodiode calibration

The slits of the monochromator were adjusted to have a wavelength band-pass of 10

nm. Because the halogen spectral maximum output was at 690 nm, the monochromator

was set at this value and the chopping factors were determined for both the reference and

test detectors using the relation

p p

lock in

V

V CF

−

− = . (3.1)

Knowing the spectral response of the reference detector, shown in Figure 3.4a, the

responsivity of the test detectors was determined by comparison using the relation

r

d

r

d

r

d

r

r lock in

d lock in

d A

A

R

R

CF

CF

V

V

R = ⋅ ⋅ ⋅ ⋅R

−

−

 (3.2)

at each wavelength increment of 10 nm, where Rr and Rd are the responsivity of the

reference detectors and test detector respectively, Vr(lock-in) and Vd(lock-in) are their output

voltages, all at the same wavelength. CFr and CFd are the chopping factors, Rr and Rd are

the feedback resistors and Ar and Ad are the sensitive areas of the reference and test

detectors respectively.22,47

The spectral response of the tested detectors is shown in Figure 3.4b. To compare

the results with the manufacturer data sheets, room temperature and manufacturer

specified bias voltage were used. The APD responsivity is directly proportional to the

wavelength of the incident light.24-27,54 This is true as indicated in Figure 3.4, for

wavelengths from 600 nm to the point where the response begins to roll off. Ideally, the

roll off point would be sharp and correspond to the energy band-gap of silicon. At this

cutoff, the responsivity decreases sharply due to insufficient energy in the incident photons
41

for the generated electrons to overcome the band-gap energy, resulting in reduction of the

APD quantum efficiency. The deviation from the ideal cutoff found in our characterized

APDs was mainly due to charge collection inefficiency of photons outside the depletion

region of the APD which was dependent on the type and level of the doping materials

used to manufacture the device.28

200 400 600 800 1000 1200 0.05

0.1

0.15

0.2

0.25

0.3

0.35

0.4

0.45

Wavelength in nm

Responsivity in A/W

600 700 800 900 1000 1100 0

10

20

30

40

50

60

70

80

90

Wavelength in nm

Responsivity in A/W

(a) (b)

RTS APD C30955E

at 317 V and 23.1 C

SLK APD C30649E

at 415 V and 23 C

o

o

Figure 3.4 Spectral response of the (a) reference detector and (b) test detectors.

3.3. Temperature dependent responsivity

At fixed bias voltage and wavelength, the responsivity of an APD detector increases

with decreasing temperature. Low temperature operation of an APD leads to an increased

output signal due to the increase in the device gain. Low temperature operation also

decreases the dark current noise level, which results in increasing the detector signal-tonoise ratio.26-27,55-58

Experiments investigating the effect of APD temperature on the spectral response

were performed. An empirical relationship for the responsivity versus temperature was
42

obtained. The nitrogen gas purge and the water circulation allowed the APD operating

temperature to be adjusted from nearly 0 oC to room temperature. It is important to

remember that a lower operating temperature causes the detector breakdown voltage to

decrease; therefore, the APD voltage bias must be chosen carefully while performing this

test to avoid device destruction.22-47

The setup shown in Figure 3.1 was used but only for the test detectors in this

experiment. During the experiment, the detector voltage bias was kept constant to insure

that the spectral response variation is only due to changes in temperature. Using the

alignment setup shown in Figure 3.3, the detector position with respect to the light source

was adjusted to the same value of the responsivity calibration experiment to insure the

same input light intensity.

The temperature controller shown in Figure 3.1 fixed the operating point for each

spectral scan. For each temperature set point, the system was allowed to reach equilibrium

before proceeding with the experiment. This whole procedure was repeated for each APD

under investigation.

For each APD the following characterization results were presented:

• The APD detector output voltage variation with wavelength, {Vn(λ)}

• The APD temperature variation with wavelength, {Tn(λ)}

Ideally, the temperature should be kept constant during this experiment. This was not true

in our case due to some deficiencies in the temperature controller used in the experimental

setup. Therefore, we had to record the temperature for each wavelength increment

{Tn(λ)}. To obtain the APD temperature, Tn, this data set was averaged according to22

T {T ( )} n = n λ . (3.3)
43

At the same temperature Tn and using the calibration data, the detector output

voltage variation was converted to a responsivity variation with respect to wavelength.

This procedure was repeated for every temperature setting giving the spectral response

variation with temperature shown in Figures 3.5a and 3.5b for the C30649E and

C30955E, respectively.

600 700 800 900 1000 1100

0

20

40

60

80

100

120

140

160

180

Wavelength in nm

Responsivity in A/W

Temperatures from top

 to bottom : 

 -3, 0, 2, 5, 8, 11, 14, 17,

 20, 23, 25 C

o

(a)

600 700 800 900 1000 1100

0

50

100

150

200

250

300

350

400

450

Wavelength in nm

Responsivity in A/W

Temperatures from top

 to bottom : 

5.7, 7.1, 9.4, 11.4, 13.9,

16.9, 20.1, 23.1C

o

(b)

5 10 15 20 25

50

100

150

200

250

300

350

400

450

Temperature in C

Responsivity in A/W

“o” at 720 nm

“+” at 820 nm

“x” at 940 nm

(d)

o -5 0 5 10 15 20 25 30

20

40

60

80

100

120

140

160

180

Temperature in C

Responsivity in A/W

“o” at 720 nm

“+” at 820 nm

“x” at 940 nm

(c)

o

Figure 3.5 APD spectral response and responsivity variation with temperature

for the (a) SLIK C30649E and (b) RTS C30955E structures respectively.
44

To obtain the responsivity variation with temperature, the analysis of the data starts

by constructing a responsivity vector, {R(T)}, and a temperature vector, {T}, at a certain

wavelength, λx, as shown in the following:

{R() ( ) ( ) ( ) ( ) T } = { } R1 λx R2 λx Λ Rn λx Λ RN λx (3.4)

{T} = { } T1 T2 Λ Tn Λ TN . (3.5)

Applying a polynomial curve fit the responsivity variation with temperature at λx, took the

form

( ) ∑= λ R = ⋅

M

m 0

m T a m T X

, (3.6)

where M is the curve fit order and N is the index for maximum temperature. The

responsivity-versus-temperature relations for the SLIK and RTS APDs are shown in

Figures 3.5c and 3.5d. The relations for the rest of the APDs are given in Appendix A.

This experiment determines the APD temperature stability requirements by obtaining

the responsivity partial derivative with respect to temperature which is given by

( ) ( ) ∑=

−

λ

= ⋅ ⋅ ∂

∂R M

m 0

m 1 m a m T

T

T

X

. (3.7)

For example, a temperature deviation (∆T) of 0.1 oC near an operating temperature of 10

o

C for the 820 nm wavelength at rated bias voltage will result in a responsivity deviation of

∆R = 0.3 and 2.2 A/W, leading to a relative error of 0.4% and 1.2% for the SLIK and

RTS structures, respectively. This indicates that the SLIK structure is more stable with

respect to temperature variation. This fact is clear from Figure 3.6 which shows the

relative error in the APD responsivity as a function of temperature at the three water

vapor absorption wavelengths of interest, assuming a 1 o

C temperature variation. Also
45

clear from this figure is that the responsivity relative error due to temperature variation

does not depend that much on the operating wavelength.47

Error curves

for SLIK APD at

 720 nm

 820 nm

 940 nm

Error curves

for RTS APD at

 720 nm

 820 nm

 940 nm

0 5 10 15 20 25

2

4

6

8

10

12

14

16

18

Temperature in C

Responsivity Relative Error in %

Figure 3.6 SLIK and RTS APD structures relative error in responsivity

due to 1 oC temperature variation obtained at 720, 820 and 940 nm

water vapor absorption wavelengths.

3.4. Bias voltage dependent responsivity

At fixed temperature and wavelength, the responsivity of an APD increases with

increasing bias voltage up to the breakdown voltage. This is due to the increased internal

electric field leading to the transfer of more energy to the photo-charge carriers which

enhances the impact ionization process. Bias voltage above breakdown causes rapid

charge carrier generation which usually leads to device damage.24-47,59

This experiment investigated the effect of the APD bias voltage variation on the

spectral response and obtained a responsivity versus bias voltage relation at the water

vapor DIAL wavelengths of interest. Similar to the previous experiment the setup shown
46

in Figure 3.1 was used for the test APDs. During the experiment, the APD temperature

was kept constant to insure that the spectral response variation is only due to the bias

voltage. The detector position with respect to the light source was adjusted to the same

value of the responsivity calibration experiment to insure the same light intensity input.

Using the high voltage supply, the detector bias is set to a certain value and the

temperature was stabilized by the temperature controller. Then, the output peak-to-peak

voltage was measured using the oscilloscope and its rms value was measured using the

lock-in amplifier to calculate the chopping factor. Next, the scan sequence started from

600 to 1100 nm with a step increment of 10 nm. For each step increment the wavelength,

output voltage and temperature were recorded using the computer. Finally, the experiment

was repeated for a set of different bias voltages. This procedure was repeated for each

APD under investigation.

At the end of this experiment, each APD had a group of data files describing its

output voltage variation with wavelength at a given bias voltage. The value of the bias

voltage, VBn, is measured directly from the high voltage supply. For a certain APD at a

given bias voltage the available data was as follows:

• The APD detector output voltage variation with wavelength, {Vn(λ)}

• The APD temperature variation with wavelength, {Tn(λ)}

The detector output voltage variation was converted to responsivity variation with respect

to wavelength using the calibration data. This procedure was repeated for every voltage

bias setting giving the spectral response variation with bias voltage.
47

To obtain the responsivity variation with bias voltage at a certain wavelength λx, we

use the spectral response variation with bias voltage to form two vectors. A responsivity

vector {R (VB)} and a voltage bias vector {VB} where defined as

{R() () () () () VB } = { } R1 λx R2 λx Λ Rn λx Λ RN λx (3.8)

B { } V1 V2 Vn VN {V } = Λ Λ (3.9)

Using a polynomial curve fit, the responsivity variation with bias voltage is given by

( ) ∑= λ R = ⋅

M

m 0

m V a m V X

, (3.10)

where M is the curve fit order and N is the index for maximum voltage bias. This analysis

was applied to each APD at 720, 820 and 940 nm wavelengths.22,47

The experimental results are shown in Figure 3.7 for the C30649E and C30955E

APDs. For each detector, the spectral response variation with bias voltage is shown on the

set of curves to the left and the responsivity variation with bias voltage on the right. The

results for the rest of the APDs are given in Appendix A.

To obtain the APD bias voltage stability, the partial derivative of the responsivity

with respect to bias voltage is obtained from

( ) ( ) ∑=

−

λ

= ⋅ ⋅ ∂

∂R M

m 0

m 1 m a m V

V

V

X

. (3.11)

For example, with both APDs operating at their rated bias voltage and for a voltage

deviation (∆V) of 1 V at 820 nm wavelength, the responsivity deviations (∆R) are 2.28

and 1.54 A/W for the SLIK and RTS APDs, respectively, which is a relative error of 4.6%

and 2% for the SLIK and RTS structures, respectively. This indicates that the RTS

structure is less sensitive to voltage variations. This fact is clear from Figure 3.8 which
48

shows the relative error in the APD responsivity as a function of the bias voltage

normalized to the APD rated bias. These relations are given at the three water vapor

absorption wavelengths of interest assuming 1-V deviation in the bias voltage. Also clear

from this figure is that the responsivity relative error due to bias voltage does not depend

significantly on the operating wavelength.47

600 700 800 900 1000 1100

0

20

40

60

80

100

120

140

160

Wavelength in nm

Responsivity in A/W

Voltage from bottomto top : 

 390, 395, 400, 405, 410,

 415, 420, 423, 425,

 427, 428, 430 V

(a)

390 395 400 405 410 415 420 425 430

0

50

100

150

Bias Voltage in V

Responsivity in A/W

“o” at 720 nm

“+” at 820 nm

“x” at 940 nm

(b)

600 700 800 900 1000 1100

20

40

60

80

100

120

140

160

Wavelength in nm

Responsivity in A/W

Voltage from bottom to top : 

310, 317, 320, 325, 330

335, 338, 340 V

(c)

310 315 320 325 330 335 340

40

60

80

100

120

140

160

Bias Voltage in V

Responsivity in A/W

“o” at 720 nm

“+” at 820 nm

“x” at 940 nm

(d)

Figure 3.7 APD spectral response and responsivity variation with bias voltage for the

(a) SLIK and (b) RTS structures respectively.
49

94 96 98 100 102 104 106 108

-2

0

2

4

6

8

10

Bias Voltage Normalized to Rated APD Bias

Responsivity Relative Error in %

Error curves

for RTS APD at

 720 nm

 820 nm

 940 nm

Error curves

for SLIK APD at

 720 nm

 820 nm

 940 nm

Figure 3.8 SLIK and RTS APD structures relative error in responsivity versus normalized

bias voltage due to 1-V deviation in the bias voltage obtained at 720, 820 and 940 nm

water vapor absorption wavelengths.

3.5. Responsivity uniformity scan

The APD sensitive area can be considered as a group of point detectors distributed

along its surface. Ideally, this distribution is uniform with each of these point detectors

having the same responsivity for similar operating conditions; therefore, the overall

responsivity distribution should be constant along the APD surface. Practically, this is not

true due to defects in the APD manufacturing process.22,28

In this experiment, we investigated the uniformity of the APD responsivity along its

surface and determined its active area. This can be achieved using a relatively small spot

size light source and scan it across the detector area measuring the APD output voltage as

a function of light spot position resulting in a responsivity map of the APD area. The small

spot size light source was achieved using the setup shown in Figure 3.9. A 633-nm He:Ne

laser was used to obtain a light beam which was focused by a microscope objective. The

position of the detector was adjusted with the computer controlled three-dimensional
50

translation stage. The motion of the detector was adjusted so that the focused laser beam

spot and the APD sensitive area remained in the same plane. A neutral density filter was

used to avoid APD saturation.

Objective

lens

APD

Detector Neutral Density

Filter

Scan

Mount Optical Axis

He:Ne

Light

Source

Figure 3.9 Responsivity uniformity scan setup.

The laser focusing optical system was calibrated to determine the displacement

between the laser focus at its minimum waist and the visual focus of the microscope

system shown in Figure 3.3. In order to obtain this calibration a pinhole was mounted on a

three-dimension translation stage, as shown in Figure 3.10. The laser focus was

determined by positioning the pinhole such that the maximum laser output was observed

on the detector. The laser focus was measured with the micrometer on the translation

stage. The visual focus was determined by viewing the best focus of the pinhole through

the eyepiece. The test detectors were positioned by finding the visual focus of the detector

surface and then translating to the laser focus by the calibrated displacement as described

above.28,60

During the scan sequence, the detector moves in a two-dimension sequence with a

fixed step size then the data is plotted and analyzed. The normalized surface scan results

for the tested APDs are shown in Figure 3.11.
51

Pin Hole

Light

Optical Axis Spot 

Laser Beam

Objective Lens

Laser Focal

Point

Figure 3.10 Laser focal point evaluation setup using a pinhole.

(a) (b)

Figure 3.11 APD surface scan for the (a) SLIK and (b) RTS APDs.

In some DIAL systems, the light intensity is focused on a small region of the

detector. Therefore, these surface scan data can be used to calculate the responsivity

correction factor since the calibration of the APD considers the average responsivity over

the entire detector active area. On the other hand, the surface scan data can be used to

determine the APD hot spots which are points on the APD surface with very high

responsivity relative to the average. The hot spots may cause a problem since at high light

intensities the charge carriers concentration will increase at this position leading to

increased power dissipation at this spot possibly resulting in local damage of the APD

area.

22
52

The areas for the SLIK and RTS APDs were measured to be 0.238 and 1.692 mm2

which is close to the manufacturer values of 0.25 and 1.5 mm2, respectively. The nonuniformity of the APD surface was defined by the ratio of the standard deviation to the

mean of the responsivity along the surface. The non-uniformity was measured to be 14.9%

and 6.1% for the SLIK and RTS APD, respectively, which indicates that this particular

RTS device is more uniform relative to the SLIK APD.

3.6. Noise measurement

The APD noise measurements were performed using a spectrum analyzer with a one

Hz normalized spectrum at 10 kHz. Appropriate care was taken to insure that the detector

dark current was the dominant noise source. The measured power spectral noise, n (in

dBm), was converted to the APD noise current spectral density, In, by the equation

f

3

L

0.1n

n R

10 R 10

I

− ⋅ ⋅ = , (3.12)

where RL is a 50 Ω APD load resistance. Table 3.1 gives the noise current spectral

density. The noise equivalent power (NEP) was obtained using the relation24-47

R= n I NEP . (3.13)

The NEP for both SLIK and RTS APDs is shown in Figure 3.12. The SLIK structure

shows a lower noise than the RTS. This might be due to the smaller detector area; thus,

the figure-of-merit (D*) was obtained for the tested detector at its responsivity calibration

bias voltage and temperature using the relation25-27,58

NEP

A D* = . (3.14)
53

This confirmed the fact that the SLIK structure has lower noise at shorter wavelengths up

to about 950 nm. The results of this experiment are very important since it directly

indicates that the SLIK would be the best detector for the water vapor DIAL detection

system based on its lower noise.47

Table 3.1 APD noise measurement results.

APD λ In NEP D*

Nm A/Hz1/2 fW/Hz1/2 cmHz1/2/W

720 1.8 2.77x1013

C30649E 820 2.3x10-13 1.7 2.90x1013

940 3.1 1.57x1013

720 28.0 4.75x1012

C30955E2 820 1.7x10-12 22.2 5.99x1012

940 19.4 6.84x1012

Figure of Merit in cmHz^(1/2)/W

600 700 800 900 1000 1100

0

2

4

6

8

10

12 x 1012

Wavelength in nm

0

10

20

30

40

50

60

70 *

o

x

+

NEP for SLIK APD

NEP for RTS APD

Figure-of-merit for SLIK APD

Figure-of-merit for RTS APD

NEP in fW/Hz^1/2

Figure 3.12 SLIK and RTS APD structures noise-equivalent-power and

figure-of-merit measurements as a function of wavelength obtained

with rated bias voltage at room temperature.
54

3.7. Frequency response

This experiment required the investigation of the responsivity variation of the SLIK

APD with respect to the frequency of the input signal and to determine its cutoff

frequency. This was done to check the manufacturer stated bandwidth and compare this

bandwidth to the whole system bandwidth according to the Nyquist criterion.51 APDs

have a very wide bandwidth in the order of giga-hertz, but due to the existence of the TIA

in the APD package, the frequency response will be limited by the internal amplifier

bandwidth which will be on the order of tens mega-hertz.24-27,39-40

The experimental setup of Figure 3.1 was modified for the APD frequency response

and shown in Figure 3.13. It consists of a 720-nm laser diode controlled by a pulse

generator. The power supply is used to bias the laser diode driver circuit consisting of a

buffer amplifier and a voltage-to-current converter. The output laser beam was split in

order to apply part of the optical signal to the reference detector while the other part was

applied to the test APD. The reference detector bandwidth was used to measure the input

signal applied to the APD under test. Next, the output of both detectors was measured by

a digital oscilloscope and a spectrum analyzer. The function of the oscilloscope was to

check the magnitude of the output signal while the spectrum analyzer measured the

frequency spectrum. When necessary, a neutral density filter was used to reduce the light

intensity to avoid saturating the test APD. The electronic high pass filter was used to

eliminate dc offsets in the detector output.22

Using the setup discussed above and after aligning the optics, we set the APD bias voltage

to its manufacturer specified value at room temperature. Then, we applied an optical

signal to the detectors after choosing a suitable neutral density filter and checked their
55

output using the oscilloscope. Next, setting the frequency of the input optical signal to a

certain value fi, by using the pulse generator the amplitude of the fundamental frequency

components of the APD input and the output signals, Ri and Ci respectively, are recorded

in dBm using the spectrum analyzer. This corresponds to one data point in the frequency

domain. By changing the frequency setting of the pulse generator and repeating the same

procedures, we obtained the complete frequency scan starting from 100 kHz to 1 MHz,

with frequency increments of 100 kHz for frequencies lower than 1 MHz, and a 1-MHz

step for frequencies up to 20 MHz. The spectrum analyzer was set to a frequency range

from 50 kHz to 20 MHz with each data point averaged 30 times.

Oscilloscope

Spectrum Analyser

Temperature

Controller

RC High

Pass Filter

Synchronizing Signal

Pulse Generator

Voltage Bias

& Power Supplies

Power Supply

Reference Detector Laser Diode Beam Spliter

Neutral

Density Filter

APD

Chamber

Input Signal

Output Signal

Figure 3.13 Frequency response experimental setup.

After collecting the data in the frequency range of interest, the analysis was done

using MATLAB.61 First, the data was stored in three vectors, {f}, {C} and {R},
56

equivalent to the frequency increments and the corresponding outputs and inputs. Then,

the gain vector {G}, in dBm, was calculated for each frequency setting from

Gi = Ci − Ri . (3.15)

The gain-frequency relation was obtained by applying a polynomial curve fit to the gain

vector with respect to the frequency vector. By plotting this relation, we can obtain the

cut-off frequency from the intersection between the curve and the -3dB line drawn from

the low frequency gain.

The previous experiment was used to obtain the frequency response for the SLIK

APD as mentioned above. The actual data and the curve fit of the frequency response are

shown in Figure 3.14. The measured cutoff frequency was 12.75 MHz which is close to

the manufacturer value (12 MHz).

3.8. Avalanche photodiode selection results

A group of APDs were characterized in order to chose a suitable detector for the

new water vapor DIAL detection system. The characterization focused on two main APD

structures, the reach through structure which is currently used in the LASE detection

system and the newer SLIK structure. The SLIK structure APD showed better

performance due to its lower noise, which will lead to an increase in the signal-to-noise

ratio of the detection system by at least a factor of 10 over the current system.17 The

EG&G SLIK APD package C30649E was selected because of the following advantages:

• Built-in, low noise, trans-impedance amplifier.

• Built-in thermoelectric cooler and thermistor for APD temperature control.
57

• The trans-impedance amplifier feedback resistance is cooled along with the

detector, reducing the Johnson noise.

105106107108 -50

-48

-46

-44

-42

-40

-38

Frequency in Hz

Input in dB

(a)

105106107108 -40

-35

-30

-25

-20

Frequency in Hz

Output in dB

(b)

105106107108 12

14

16

18

20

22

24

Frequency in Hz

Gain in dB

(c)

Figure 3.14 SLIK APD (a) input and (b) output variation with frequency and (c)

frequency response obtained at 336V bias and 23 oC temperature.

A drawback in the selected APD arises from its small area (0.24 mm2

) compared to

the LASE APD (1.7 mm2

) which might produce alignment difficulties. However, its lower

noise strongly recommends it for the development of the new system.
58

CHAPTER IV

DETECTION SYSTEM DESIGN

AND LABORATORY PERFORMANCE

The main goal of this research effort is to increase the signal-to-noise ratio (SNR) of

the water vapor DIAL detection system by a significant factor compared to the LASE

instrument.7,18 Also, it is required to design the system to a compact size suitable for

mounting directly on the receiver telescope with output data compatible with a simple

computer interface. In this chapter, the design details of the water vapor DIAL detection

system will be discussed, as well as its laboratory performance.18-20,62

4.1. Detection system design

A block diagram of the new water vapor DIAL detection system is shown in Figure

1.2. The APD detected light signal is applied to the analog circuit shown in the block

diagram of Figure 4.1. The analog circuit is designed to control the operation of the APD

package and to condition its output signal in preparation for the 14-bit digitizer.

The laser return signal is focused onto the 0.5-mm diameter APD sensitive area. The

APD output signal is applied to the signal conditioning stage, which consists of three

substages. In DIAL applications, the transmitted laser pulses might hit a large amount of

water vapor or aerosol layer, such as clouds, resulting in high levels of backscattered light

which could overload the system. These overload signals are a major problem since they

saturate the detector and cause “hanging” of the digitizer, which leads to loss of some data

words. The clipping circuit is used to protect the digitizer by clipping these overload
59

signals to a predefined level. For normal signals the clipping circuit acts as a voltagefollower amplifier. The APD overload is prevented using a current limiting resistor R as

shown in Figure 4.1.62

High Voltage

Bias Cuircuit

Bias

Controller

Temperature

Conroller

Clipping

Circuit APD Bessell Filter

Detected

Signal

Analog

Signal

Optical

Signal

Summer

&

Buffer Amplifier

Dither Input

Marker Input

R Signal Conditioning Stage

Figure 4.1 Block diagram of the analog circuit.

Following this stage is a summer and buffer amplifier, designed to provide an

additional gain to the detected signal in order to achieve a 2-V maximum peak-to-peak

value which is compatible with the digitizer maximum input range. Also, it adds a marker

and dither signals to the detected signal. The marker signal is used to mark the beginning

of the useful data, while the dither is used to add a low level sinusoidal signal for better

digitization performance.18,62

Finally, the signal is applied to a Bessell filter, with a 2.5-MHz cutoff frequency. This

filter limits the system bandwidth in order to reduce the high frequency noise and restrict

the signal frequency with respect to the digitizing frequency according to the Nyquist

criterion.39,51
60

Since the APD responsivity is a strong function of its voltage bias and temperature,

two proportional integral (PI) controllers were used. The first is a voltage controller which

can be adjusted manually to apply a constant bias to the APD. The second is a temperature

controller with a fixed set point. This controller is used to cool the detector with respect

to the ambient temperature.68

The output, detected signal from the analog circuit is applied to the digital circuit.

The digital circuit was designed mainly to operate as a waveform digitizer but also

performs some simple data processing such as averaging DIAL lidar return signals

(hardware averaging) and monitoring the detection system performance. Figure 4.2 shows

a block diagram of the main components of the digital circuit. The 14-bit, 10-MHz analogto-digital converter (ADC) is used to convert the analog lidar signal from the detector to a

digital format. The dual-port RAM is used for temporary data storage and it also isolates

the 10-MHz digitizer frequency from the 16-MHz microcontroller frequency.19-20,62

14 Bit

16 Bit 8 Bit

16 Bit 8 Bit

Trigger Input

14 Bit, 10 MHz

Analog to Digital

Converter

16 Bit, 16k

Dual Port

Static RAM

16 Bit, 16 MHz

Microcontroller

2x9 Bit, 8k

FIFO Analog

Signal To Computer

Interface

16 Bit

Clock Input

Figure 4.2 Block diagram of the digital circuit.

Approximately 400 μs before the DIAL laser fires, a trigger pulse is generated and is

used as a precursor that informs other systems that the laser is about to fire. This pulse is

sensed by the microcontroller and acts to synchronize the detection system with the laser

transmitter. A 10-Hz maximum laser firing frequency is assumed in this design, although a
61

higher repetition rate could be accommodated. Also, the microcontroller will be used for

on-line data averaging and house keeping.20

Finally, the first-in-first-out (FIFO) memory is used to output the final data to the

recording system. Again, the FIFO separates the 16-MHz microcontroller operating

frequency from the output data transfer rate which is dependent on the computer interface.

The computer interface used with the detection system was a parallel input-output

interface with a maximum of 20 MSPS data transfer rate. This interface is fully

controllable by C++ software. The final data is stored in an output file which is accessible

with any other software package.

4.1.1. Analog circuit

The signal conditioning stage is the most critical stage in the whole detection system

since any noise contribution from this stage to the detected signal will directly affect the

useful data. Figure 4.3 shows the circuit diagram of this stage. In the text the various

components of the system are referred to by their letter designation. Details of circuit

schematics are given in Appendix B, and the component listing is given in Appendix C.

Operational amplifier U7 is a clipping op amp which operates as a voltage follower for the

detected signal provided that the signal lies within the clipping limits. The upper clipping

limit is set to 2.5 V using R24 and R25. This limit is chosen only for the op amp stability

and has no influence on the output since the detected signal coming from the APD

package is always negative with a maximum theoretical value of zero. The lower clipping

limit is set by a −1.5 V reference voltage (Appendix B). Resistor R26 acts as the load

resistance for the op amp as recommended by the manufacturer. The output of the
62

clipping op amp is then applied to U8 which is a gain and summer amplifier. Using this op

amp the signal gain is set to 3 using R27 and R26. Another gain of 0.5 is applied to the

signal using the R30 and R31 voltage divider, leading to a total end-to-end gain of 1.5

which was chosen so that the maximum APD output fit the maximum digitizer limit. This

voltage divider is used for termination according to the maximum power transfer condition

of Thevenin’s theorem.51 The summation of the marker and the dither inputs to the

detected signal is obtained using R28 and R29, respectively.40,52, 63,64

+5V

+5V

-12V

-5V

+5V

-5V

+5V

-1.5V Reference

SIGNAL

HIGH VOLTAGE

THERMISTOR 1

THERMISTOR 2

TEC

R23

140

R25

10k

JP3

R31

5 0

R27

300

R28

100

R29

100

R26

100

+

-

U 7

AD8036

3

2

6

78

45

R22

130

C16

0.1uF

C19

0.1uF

R30

5 0

JP4

U 6

APD

7

1 1

6

9

1 2

8

2 1

1 0

3

5

4

+

-

U 8

AD8041

3

2

6

7 4

R21

5 0

R24

10k

U 9

B3LL31-2.5

Input GND

GND Output

Marker

Dither

 Check

Figure 4.3 Circuit diagram of the signal conditioning stage.

The final output is then filtered using U9 which is a 3 pole, 2.5-MHz, low-pass

Bessell filter. The Bessell filter was selected due to its excellent performance regarding the

phase shift for real time data. In order to check the output signal from this stage, the
63

voltage across R31 can be accessed using an external connector. Also, this connector can

be used to inject an external voltage to the digitizer according to the JP3 and JP4 jumper

settings.65

Figure 4.4 shows the circuit diagram of the APD high voltage bias controller. The

APD voltage bias is supplied from U4, a high voltage module. The control of the high

voltage was obtained using U3, a dual op amp chip, with one op amp acting as the

proportional controller set to a gain of 4 using R3 and R4 and the other as the integral

controller set to an RC time constant of 0.04 s using R5 and C5. Diode D1 is used to

ensure positive control voltage to U4. Potentiometer R7 is used to set the APD high

voltage between 292 and 365 V. The output high voltage is applied to a low-pass filter

formed by R8 and C8 to eliminate any bias ripples. In addition, R8 limits the APD current

for overload protection, and C9 supplies it with instantaneous in-rush current. A potential

divider formed by R9 through R13 is used as the voltage feedback to the controller.

Voltage monitoring is obtained after the proportional controller with a voltage reading

VBM related to the high voltage bias VBIAS by the equation66,67,51

20

14.91

V V BIAS

BM = − . (4.1)

The APD package was supplied from the manufacturer with a built in TEC cooler

and a thermistor. The temperature status of the detector is sensed by the thermistor by

placing it in an arm of a Wheatstone bridge formed by R15, R16, R17 and R18, as shown
64

+12V

+12V

-12V

HIGH VOLTAGE

Bias Monitor

+5V Reference

D1

1N4148

C5

2uF

R6

500

R5

200k

+

-

U3B

AD706

5

6

7

R4

200k

R7

10k

+

-

U3A

AD706

3

2

1

8 4

R10

2.55M

R11

2.55M

R12

82k

R13

22k

R9

2.55M

R3

50k U4

521-5-M

+ 5V Ref

Power

Control

HV out

GND

GND

C9

 22pF

R8

10k

C8

0.1uF

Bias Monitor

Figure 4.4 APD voltage bias controller circuit diagram.

in the circuit diagram of Figure 4.5. The value of R18 determines the balance condition of

the bridge therefore setting the detector temperature. Since the thermistor resistance RT is

given by49





 



 



 



 −

= ⋅ 298

1

T

1 3940

4

T R , (4.2) 10 e

and the bridge balance condition is51

1

R17 R18

R

R16

R15 T = + = . (4.3)

Therefore, the temperature setting, in Kelvin, will be given by

1

4

3

298

1

3940

10

R18 33x10 ln

T

−





























+















 +

= . (4.4)

Equation (4.4) indicates that the minimum temperature setting is selected by R17 (33 kΩ)

and is equal to 0.3 oC with R18 short-circuited. Temperatures lower than this value usually

cause problems due to condensation of water vapor on the detector window. Zener diode
65

D2 is used to supply the bridge by its zener voltage VZ equal to 5.1 V. The zener current

limit is set by R14.53

+5V

+12V

+12V

THERMISTOR 1

THERMISTOR 2

Temperature Monitor

TEC

R32

2.5,10W

R16

33k

R15

33k

R17

33k

R14

2 k

R20

10k

D 2

1N751

C10

1uF

R19

200k

Q 1

TIP110

R18

TBD

+

-

U5B

AD706

5

6

7

+

-

U5A

AD706

3

2

1

8 4 Temperature Monitor

Figure 4.5 APD temperature controller circuit diagram.

The bridge balance is sensed by an instrumentation amplifier formed by the dual op

amp U5,52 with one op amp acting as a voltage follower for the temperature monitor and

the other acting as both the proportional integral controller with RC time constant of 0.02

s set by R19 and C10. The controller output is applied to a potentiometer R20 to set the

TEC current by controlling the collector emitter current of the Darlington transistor Q1.68

Resistor R32 is a 10 W resistor used to set the maximum TEC current to 2 A according to

its manufacturer specification. The temperature monitor voltage reading VTM will be given

in terms of the zener voltage by53

Z

T

TM V

R17 R18 R

R17 R18 V

+ +

+ = . (4.5)
66

After starting the circuit when the detector temperature reaches its steady state value the

monitor voltage will be simply given by

TM VZ 2

1 V = . (4.6)

This voltage can be read directly using an output connector to indicate the temperature

stability of the APD.53

Referring to Figure 4.3, the conditioned analog signal “SIGNAL” will be applied to

the digitizer, provided that both jumpers are shorted. A high input impedance instrument,

such as an oscilloscope, can access this signal using the “CHECK” terminal.

4.1.2. Digital circuit

The conditioned analog signal was applied to a 14-bit, 10-MHz ADC, as shown in

the block diagram of Figure 4.6. The input digitizer range was set to be from 0 to 2 V.

Therefore, the maximum allowable output signal from the detector package is given by

1.33V

3

2 2

Signal Conditioning Gain

ADC Upper Limit 2 = − − × = × (4.7)

while the clipping op amp U7 lower limit was set to −1.5 V. This was done to insure the

linearity of the signal conditioning stage by avoiding the non-linear knee between the linear

and clip regions in the op amp input-output characteristics.

A 10-MHz clock signal was externally applied to the ADC. Therefore, it

continuously digitizes the analog signal during each clock cycle. The digitized data was

buffered and stored in a certain memory location in a 16-bit, 16 k, dual-port RAM. The

data storage location or address was set by a 13-bit counter formed by cascading three 4-

bit binary counters and a JK flip flop. The counter and the RAM share the same 10-MHz
67

clock in order to synchronize them with the ADC. Connecting the most significant bit

(MSB) of the left RAM address to the supply enables the counter only to access its upper

half while the lower half of the RAM is kept for data averaging proposes. Thus, the

maximum real time window for the digitized data can be obtained by

819.2 sec

10 10

16 1024 2

Clock Frequency

StorageSpace Data Time Window = 6 = μ

×

× = (4.8)

which is enough to record both the on and off line DIAL laser signal returns.

13 Bit Counter

ADC

Left RAM

Buffer

OTR

BIT 1-12

A13

A0-12

I/O0-12

Start Counter

End of count

Clock input

ENABLE

CLK

LOAD Clear Counter

H

R/W

CLK

13 BIT

12 BIT 13 BIT

Figure 4.6 Block diagram of the data collect and store circuit.

The counter operation is controlled by the microcontroller. At the beginning of the

circuit operation, the microcontroller sends a clear signal to the counter resetting its

output to the left address 2000H equivalent to the right address C000H which is the first

storage location in the RAM, as shown in Figure 4.7. The ADC output will be

continuously stored in this location until the trigger signal is received.
68

3FFFH FFFFH

ADC Output

Data Storage

2000H C000H

1FFFH BFFFH

Data Averaging

0000H 8000H

7FFFH

FIFO

7000H

6FFFH

Control Word

6000H

5FFFH

NOT USED

4000H

3FFFH

Internal Program

Storage EPROM

2080H

NOT USED

00FFH

Register File

0000H

Figure 4.7 System memory map.

When the microcontroller senses the trigger signal, it sends a start count or “counter

enable” signal to the counter causing it to address successive memory locations in the

RAM in order to store the useful data. After scanning half the memory, when the counter

reaches its maximum count of 1FFFH, which is equivalent to memory in the address of

3FFFH, an “end-of-count” signal will be sent back to the microcontroller indicating the

end of the time window record. Therefore, the useful data will be available to the

microcontroller in the address range from C000H to FFFFH, as shown in the memory

map in Figure 4.7.69-73
69

The 16-MHz, 87C196KB-16 microcontroller is shown in the block diagram of

Figure 4.8. The memory map of the microcontroller and the RAM is shown in Figure 4.7.

The microcontroller address and data are shared on the same bus using address latching.

The microcontroller address is decoded to access the right address of the RAM, a FIFO

and a dip switch. The memory section, starting with the nibble 7, is reserved to access the

FIFO as will be discussed in the output stage. While the memory section, starting with the

nibble 6, was reserved for the control word, set by on-board 8-bit dip switches S1, the

“not used” address range from 4000H to 5FFFH can be used for system upgrades to the

87C196KC microcontroller.74-78

After storing the ADC data in the RAM, the counter sends an “end-of-count” signal

to the microcontroller. The microcontroller starts to transfer and add the data from the

upper half of the RAM to its lower half for averaging. At the end of each averaging cycle,

the data will be transferred to the FIFO which is waiting to transfer data to the reading

device. The dip-switch output is buffered and was used to give the microcontroller a

control word to indicate the averaging times required with a maximum of 4 averages.

Also, it indicates the record length which could be 1, 2, 4 or 8 k words corresponding to

102.4, 204.8, 409.6 or 819.2 μs, respectively. (In case of record length less than 8k, the

ADC data will still be stored in 8k and the first 4k, 2k or 1k will be considered and the rest

of the data will be neglected.)62

One advantage of the selected microcontroller is its built-in 8 channel 10-bit ADC,

two channels of which are reserved for the detector bias voltage and temperature monitor

readings.75-76
70

Right RAM

A0-13

I/O0-15

Microcontroller

Address Latch

AD1-14

AD0-15

D0-15

13 BIT

13 BIT

13 BIT

Control Word

DIP Switch Decoder

FIFO Decoder FIFO Enable

Figure 4.8 Microcontroller connection block diagram.

Figure 4.9 shows the block diagram of the output stage of the digital circuit. The

FIFO is formed by a parallel connection of two 8-k, 9-bit FIFO’s memories to form an 18-

bit word to handle the 16-bit data.77 The output data was buffered and applied to four 4-

channel, TTL-compatible optical isolators. These isolators are used to isolate the circuit

and the reading device grounds.79 Finally, the data was read using a SCXI connector. The

FIFO write and reset operations are controlled by the microcontroller, while the FIFO

read is controlled by the reading device.80 In order to check the correct sequence of data

handling, the empty flags of both FIFOs are read by the microcontroller. Any unequal state

of these flags indicates a FIFO failure or data crash.

After the microcontroller sends the complete data set to the FIFO, it sends a “FIFO

ready” signal to the reading device to synchronize it with the system. Both the FIFO ready

and FIFO read signals are also optically isolated by U37 which are opto-couplers with

dual separate channels, once again to separate the grounds to minimize noise pick up.
71

Finally, the microcontroller controls the FIFO reset signal in order to reset the FIFO

before the first writing operation.62

Microcontroller Isolation

AD0-15

D0-15

16 BIT

Data Ready

FIFO

2xD0-7 2xQ0-7

FIFO Read

Port 1.0

Buffer

16 BIT 16 BIT

R

Figure 4.9 Block diagram of the output stage.

4.2. Detection system settings

This detection system was built on one 312-gm, 15x9 cm2 printed circuit board using

surface-mount technology. This was very challenging because of the sensitivity of the

different elements to the board layout, since analog signals were subjected to different

noise pick-up sources from the high switching frequency digital signals. The board layout

was carefully designed to separate the analog and digital circuits on the two board sides

with ground isolation between them. The whole detection system is optically isolated from

any external instrument such as the clock generator, trigger input and computer. The only

electrical connection between the detection system and any external device will be through

the power supplies that had proper filtering and isolating the power grounds from the

circuit grounds. The calculated power consumption of the card is 21.1 W (see Appendix

C).18-20,62
72

4.2.1. Microcontroller program

Figure 4.10 shows the flowchart of the microcontroller program. After switching on

the power supplies, the start sequence begins by resetting the microcontroller chip and

configuring its port, as shown in Figure 4.11. Port 1 is used for the counter and FIFO

control and monitoring, while port 2 is used for sensing the trigger input and serial

communication if used.74

The microcontroller then reads the control word from the dip switch for averaging

and record length settings, as shown in Figure 4.12. Then, it starts an infinite loop by

clearing the counter in order to address the first data storage location in the RAM (2000H

as shown in Figure 4.7). The microcontroller then enters a waiting loop for the trigger

input. This trigger comes from the lidar transmitting system. If the trigger signal is sensed,

the microcontroller starts the counter in order to store the output digitized data from the

ADC. During this time, the microcontroller waits for the “end-of-count” signal from the

counter.

If the microcontroller sensed the end of count signal, indicating the storage of the

useful data, it starts transferring the data. The amount of the transferred data is 1, 2, 4 or 8

k words depending on the record length setting. If averaging was not used, the data will

be transferred directly to the FIFO, and the data ready signal will be transmitted to the

reading device. The sequence will be repeated in the next cycle. If averaging was used, the

data will be transferred to the averaging memory locations as indicated in Figure 4.7; then,

the cycle will be repeated to acquire additional data to be added to the previous data.

After completing the number of averages, the final averaged data will be sent to the FIFO,

and a ready signal will be transmitted to the reading device and the sequence repeated.
73

Start

Read DIP Switch

Clear Counter

Trigger

?

Start Counter

End of Count

?

Average = 1

?

Transfer/Add

Data

End of

Average

Transfer to FIFO ?

Data Ready

YES NO

YES

NO

YES

NO

YES

NO

Figure 4.10 Flowchart of the microcontroller program.

P2.7 P2.6 P2.5 P2.4 P2.3 P2.2 P2.1 P2.0

TRG SO SI

Input / Output Port 2

P1.7 P1.6 P1.5 P1.4 P1.3 P1.2 P1.1 P1.0

CC CE EOC RF UBEF LBEF FR

Input / Output Port 1

RF

FR

LBEF

UBEF

EOC

CE 

CC

: Reset FIFO

: FIFO is Ready

: Lower Byte Empty Flag

: Upper Byte Empty Flag

: End Of Counting

: Counter Enable

: Clear Counter

SI

SO

TRG

: Serial Communication Input

: Serial Communication Output

: Trigger Signal

Figure 4.11 Microcontroller ports 1 and 2 configuration.
74

DS1 DS2 DS3 DS4 DS5 DS6 DS7 DS8

8 Bit Dip Switch (S1)

ON = 0

OFF = 1

ON

ON

Record Length Average

DS7 DS8 Average

1 1

1 0

0

0 1

0 None

2

3

4

DS5 DS6 Record Length

1 1

1 0

0

0 1

0 1 k Words

2 k Words

4 k Words

8 k Words

TBD TBD TBD TBD

Figure 4.12 Dip switch configuration.

4.2.2. Data acquisition card

The final data stored in the FIFO is transferred to a personal computer using the

PCI-DIO-32HS parallel input-output data acquisition card manufactured by National

Instruments, Inc,.80 The PCI-DIO-32HS is the state-of-the-art input-output data

acquisition card which can be software configured using C++ language and can acquire

data with up to a 20-MHz data transfer rate.81 The flowchart of the card program is shown

in Figure 4.13 while the card connection to detection system is shown in Figure 4.14.

The card waits for the microcontroller data ready signal which is sensed by an

acknowledge pin (ACK). If this signal is set, the card will start a pattern of data request

signals (REQ) which will be used to read the FIFO using two parallel ports. This data set

will be stored in a vector. Successive data sets will be added to this vector according to

the number of the software averaging setting. The final averaged data is stored in an

output file representing the DIAL detected signal.
75

Send REQ

Average

?

Store Data

Acquire Data

YES

NO

Data Ready

?

YES

NO

Record Length

?

YES

NO

Output Data File

Start

Figure 4.13 Flowchart of the PCI-DIO-32HS data acquisition card. DIAL Detection System PCI-DIO-32HS

Data Ready

FIFO Read

Data

REQ

ACK

Input Ports

Figure 4.14 PCI-DIO-32HS data acquisition card connection to the detection system.
76

The whole system operation is summarized in the data flow diagram of Figure 4.15.

The limitations of the averaging time TA, transfer to FIFO time, TF, and the external

transfer time, TE, are given in table 4.1.

Transfer to FIFO External Transfer

Signal Processing Data Transfer

TF TE

Digitization

Trigger

819 2. μs

Averaging

TA

Figure 4.15 Detection system data flow diagram.

Table 4.1 Timing intervals for different record lengths.

Record Length TA TF TE Laser Frequency

k word μs Ms ms ms Hz (maximum)

1 k 102.4 10.3 11 0.6 90.1

2 k 204.8 20.5 22 1.2 45

4 k 409.6 41 44 2.5 22.5

8 k 819.2 82 88 4.9 11.3

4.3. Laboratory performance testing

The laboratory performance testing was performed on the individual circuits and for

the whole system. The setup of the detection system performance testing is shown in

Figure 4.16. The DIAL return signal was simulated using a 788-nm wavelength laser

diode. A diffuser and neutral density filter (NDF) with variable settings were used to
77

condition the optical signal in terms of uniformity and intensity. The check point was used

to either observe the analog circuit output or to inject an external signal to the digital

circuit.

Oscilloscope

Pulse Generator

PCI-DIO-32HS

Interface

Personal

Computer

DIAL

Detection

System

Optical

Axis

Laser

Diode

Diffuser

APD

Power Supply

NDF

Pulse Generator

Pulse Generator

CLK

TRG

Delay

Data

Check TM BM

Figure 4.16 Experimental setup for detection system laboratory performance testing.

4.3.1. Temperature and bias voltage controllers performance

The APD temperature was set to 0.3 o

C by setting R18 equal to zero. For an initial

APD temperature of 23.6 o

C, Figure 4.17 shows the transient response of the APD

temperature monitor voltage VTM, the thermistor resistance, the APD temperature and the

TEC current. The rise time is approximately 5 s, and the 5-percent settling time is

approximately 50 s with about 2 oC of overshoot. The steady-state temperature was

0.3±0.3 oC. The temperature measurement is obtained by converting the temperature

monitor voltage to a thermistor resistance according to equation (4.5), and then this is
78

further converted into temperature according to equation (4.4). From Figure 4.17a and c,

it is interesting to note a linear relationship between the APD temperature, T, and the

temperature monitor voltage, VTM over a 30 oC operating range which is approximated by

T 17.2 V 43.6 ≈ ⋅ TM − . (4.9)

(c) APD Temperature Variation

(a) Temperature Monitor Output (b) Thermistor Variation

(d) TEC Current

-20 0 20 40 60 80

2.2

2.4

2.6

2.8

3

3.2

3.4

3.6

3.8

4

Seconds

Volts

0 20 40 60 80 100

-0.05

0

0.05

0.1

0.15

0.2

0.25

0.3

0.35

0.4

Seconds

Amps

-20 0 20 40 60 80

10

15

20

25

30

35

40

Seconds

k Ohms

-20 0 20 40 60 80

-5

0

5

10

15

20

25

 Seconds

C

o

Figure 4.17 APD temperature controller response (a) temperature monitor, (b) thermistor

resistance variation, (c) temperature variation and (d) TEC current.

Figure 4.17d shows a steady-state TEC current of approximately 0.35 A which

corresponds to a steady state TEC input power of 174 mW. For successful temperature
79

controller operation, the maximum ambient temperature is 27 oC. Since the controller was

designed to cool the detector, the minimum ambient temperature is theoretically equal to

the controller temperature setting.53

The APD voltage bias controller was set to produce a bias voltage of 336 V. This

value was chosen to avoid breakdown at low operating temperature. Figure 4.18 shows

the turn-on transients APD voltage bias monitor and its bias voltage obtained using

equation (4.1). The measured steady state bias is 336.6±0.6 V with a settling time of 1.3 s

and rise time of 75 ms with an undershoot of 297.4 V.

-2 0 2 4 6 8 10 12 14

295

300

305

310

315

320

325

330

335

340

second

Volt

-2 0 2 4 6 8 10 12 14

-0.5

0

0.5

1

1.5

2

2.5

second

Volt

(a) Bias voltage monitor transient. (b) APD bias voltage transient.

Figure 4.18 APD voltage bias monitor and bias voltage transients.

4.3.2. Signal conditioning circuit performance

In order to evaluate the signal conditioning stage gain, the APD output was replaced

by a calibrated voltage input, and the output was then measured. Figure 4.19 shows the

normalized gain versus the input voltage. The mean gain along the input range of the

signal conditioning stage was measured to be 1.52 with 0.2 % standard deviation. The
80

clipping action of this stage is clear and starts at an input signal of 1.33 V corresponding

to an output of 1.99 V which is compatible with the ADC. The frequency response of the

signal conditioning stage is shown in Figure 4.20. This was measured by applying a known

sinusoidal input and measuring the change in the amplitude and phase due to the change in

the frequency. The input frequency was varied from 1 kHz to 3 MHz. The cutoff

frequency at -3 dB is 2.4 MHz which is compatible with the digitization frequency of 10

MHz.82

0 200 400 600 800 1000 1200 1400 1600

0.95

0.96

0.97

0.98

0.99

1

1.01

1.02

1.03

1.04

1.05

Input voltage in mV

Normalized Gain = (Vout/Vin)/(mean gain)

Figure 4.19 Signal conditioning stage normalized gain versus input voltage.

Figure 4.21 shows the APD and the signal conditioning stage noise spectrum

obtained with no input (dark conditions) using a spectrum analyzer with 400 averages.

This measurement suggested the use of batteries instead of power supplies to eliminate the

power frequency harmonics noise pickup shown in the figure. The battery operation
81

spectrum shows the 1/f noise with a knee at 3 kHz. Pickup noise at the fundamental power

frequency is clear from the figure and can be eliminated by proper shielding in the final

packaging. The average integrated noise was found to be 150 nV/Hz1/2 in the operating

frequency range which indicates that the APD noise contribution is the dominant source.

Considering the 2.5 MHz system bandwidth and the APD responsivity, this corresponds to

a minimum detectable signal level of 1.5x10-5 W/m2

.

103 104 105 106 107

0.6

0.8

1

Normalized Gain

Frequency in Hz

103 104 105 106 107

-400

-300

-200

Phase Shift

Frequency in Hz

(a)

(b)

Figure 4.20 Signal conditioning stage frequency response

for the (a) gain and (b) phase shift.

4.3.3. Digitizer performance

Several operational tests were performed to check the operation of the digital

circuit. These tests included the following:

• The microcontroller ability to access the dip-switch, RAM and FIFO.
82

• The counter ability to address the RAM.

• The interface card ability to read the FIFO.

102103104105 10-7

10-6

10-5

10-4

Frequency in Hz

Noise in nV/Hz ^1/2

102103104105 10-7

10-6

10-5

10-4

Frequency in Hz

Noise in nV/Hz ^1/2

(a)

(b)

Figure 4.21 APD and signal conditioning stage low frequency noise spectrum

using (a) power supplies and (b) batteries.

In order to test the performance of the whole digitizer circuit, a histogram test was

applied to check for any missing codes. In the histogram test, a sine-wave is applied to the

digitizer input with a non-coherent frequency with the sampling rate of the digitizer. The

amplitudes of this signal must be selected slightly out of the digitizer range to insure the

coverage of all possible codes. Several samples of the input signal were taken and the

occurrence of every possible ADC output is counted.83 The selected sine-wave for this test

had a 4.95-kHz frequency with 2.03 V peak-to-peak and 997.3 mV offset levels, shown in

Figure 4.22a. The over and under range data shown in the figure insures that the input

signal covers the whole digitizer range; therefore, all codes must be present in the
83

histogram plot. The histogram plot is shown in Figure 4.22b for 500 cycles. The height of

each point is proportional to the total number of times that code occurred. Missing codes

would appear at zero height. Figure 4.22b indicates no missing codes.

0 2000 4000 6000 8000 10000 12000 14000 16000

0

2000

4000

6000

8000

10000

12000

Counts

Number of occurrence

(b) Histogram plot

0 1000 2000 3000 4000 5000 6000 7000 8000

0

0.5

1

1.5

2

2.5

3

x 104

Samples

Counts

Over range = 32767

Under range = 16384

(a) Digitizer output

Figure 4.22 (a) Input sine-wave signal to the digitizer and (b) the histogram

plot which indicates no missing codes.

4.3.4. Overall system performance

Testing of the whole detection system was performed in order to characterize its

gain and SNR. The input optical signal was obtained from a 788-nm laser diode

corresponding to an APD responsivity of 117.5 A/W at a bias voltage of 336 V and 0.3 o

C

operating temperature. The laser diode current and temperature were kept constant using

a driver circuit and a temperature controller to stabilize its operation. Single-shot pulsed

signals with 100 μs duration and variable intensities were obtained to measure the gain and

SNR. Neutral density filters were used for changing the laser input intensity. Using the

setup shown in Figure 4.16, the experimental intensity level was varied between a
84

minimum of 8.5x10-5 W/m2 to a maximum of 8.42x10-2 W/m2 to insure the linearity of the

system. Figure 4.23a shows the system output in counts versus the input light intensity

which indicates a linear relation with a constant total gain of 183,960 counts/(W/m2) over

the characterized range. Figure 4.23b shows the SNR versus the input laser intensity. The

SNR was obtained by squaring the ratio of the detected pulse mean value to its standard

deviation. A minimum SNR of 2.7 was obtained in this experiment.

10-4 10-3 10-2 10-1

0

5000

10000

15000

Output in Counts

Gain = 183960 Counts W m ( )2

Intensity m (W ) 2

SNR

10-4 10-3 10-2 10-1

0

1

2

3

4

5 x103

Intensity m (W ) 2

(a) (b)

Figure 4.23 (a) System output in counts versus input light intensity obtained with no

averaging at 788 nm and (b) the corresponding signal-to-noise ratio.

In order to determine the minimum measurement capabilities of the system, the lidar

return signals were simulated in order to obtain the minimum water vapor detection

capabilities of the system. To convert the lidar detected signals into a water vapor number

density profile the DIAL equation is used which is given by

( ) () ()

() () N R

R

P RPR

P RPR

OFF ON

OFF ON

= ⋅ ⋅

⋅

⋅

















1

2

2 1

1 2 ∆ ∆σ ln , (4.10)
85

where N is the water vapor number density, ∆R is the range cell, ∆σ is the differential

absorption cross section and PON and POFF are the DIAL on-line and off-line return signals,

respectively. Ideally, if the on-line and off-line return signals are identical, the water vapor

number density should be zero, as indicated in the above equation.

Applying this concept to the detection system, one can estimate the minimum

detectable water vapor number density, which is determined by the system noise as shown

in Figure 4.24. Therefore, the laser diode was configured to give two identical exponential

decaying pulses with a 30 μs time constant as shown in Figure 4.24a. The first pulse was

to simulate the on-line return while the second pulse was to simulate the off-line pulse.

Figures 4.24b and c show the single shot water vapor number density obtained for

the lower and upper troposphere, respectively, with no averaging. The DIAL calculations

were performed using a range cell of 300 m and water vapor differential absorption cross

section of 10x10-24

 cm

2

 and 150x10-24 cm

2

 for the lower and upper troposphere,

respectively, as specified by the LASE detection system. The DIAL calculation indicated a

minimum detectable water vapor number density of 6.7x1016 cm-3 for the lower

troposphere and 4.5x1015 cm-3 for the upper troposphere. These water vapor number

densities are comparable to those for the LASE instrument which are 5x1017 cm-3 and

5x1014 cm-3 for lower and upper troposphere, respectively, with 1000 shot averages. This

indicates the improved water vapor detection capabilities of the new system by a factor of

9 for the upper troposphere measurements.20
86

0 1000 2000 3000 4000 5000 6000 7000 0

2000

4000

6000

8000

10000

12000

14000

Water Vapor Number Density ( cm ) -3

Samples

Output in Counts

∆n = × 4 5 1015 . cm -3

0 10 20 30 40

-1

-0.5

0

0.5

1

x 1016

Altitude ( km )

0 10 20 30 40

-1

-0.5

0

0.5

1

x 1017

Altitude ( km )

∆n = × 6 7 1016 . cm-3

On-line Off-line

(a)

(b)

(c)

Figure 4.24 (a) Simulated lidar return signals. (b) Water vapor number density noise floor

for the lower troposphere and (c) for the upper troposphere.
87

CHAPTER V

THE LIDAR THEORY

In this chapter, the derivation of the backscattering lidar equation will be discussed.

The modification of the equation to include the absorption and the geometry of the

receiver system also will be given. Based on the backscattering-absorption lidar equation,

the differential absorption lidar (DIAL) equation will be derived. Finally, application of the

backscattering lidar equation for measuring aerosols will be discussed as well as the

application of the DIAL equation for water vapor measurement.

5.1. The Lidar Equation

The scattering form of the lidar equation defines the received atmospheric

backscattered laser power in terms of laser pulses propagated into the atmosphere and

collected by the receiver that is located near the laser transmitter. The variation in the

received intensity is used to determine the density profile of atmospheric molecules and

aerosols as a function of the altitude.

Under the assumption of a pulsed lidar, the variation in the signal power ∆P(λ,R)

received by the detection system in the wavelength range (λ,λ+∆λ) from the range

element (R,R+∆R) is given by5

( ) ( ) ( )( ) ∫ ∆P , (5.1) λ,R = J λ,R,r ⋅∆λ ⋅∆R ⋅ p λ,R,r dA R,r

where J(λ,R,r) is the laser-induced spectral radiance at wavelength λ, at position r in the

range R per unit range interval, p(λ,R,r) is the probability that radiation of wavelength λ
88

scattered from position r at range R will be detected and dA(R,r) is the element of the

target area at position r and range R.

The target spectral radiance J(λ,R,r) depends on the nature of the interaction

between the laser radiation and the target medium. For elastic or inelastic scattering

medium, J(λ,R,r) will be given by5

J( ) ( )( ) ,R,r , ,R,r I R,r L λ = β λ λ ⋅ , (5.2)

where I(R,r) is the laser intensity at position r and range R and β(λL,λ,R,r) is the volume

backscattering coefficient given by5

( ) () ( ) ∑ ⋅I ( ) λ 











Ω

σ λ β λ λ = ⋅ i

i

s

i

L

L i d

d , ,R,r N R,r , (5.3)

where Ni(R,r) is the number density of scatterer species i, {dσ(λL)/dΩ} is the differential

scattering cross section under irradiation with laser radiation at wavelength λL and Ii(λ) is

the fraction of the scattered radiation that falls into the wavelength interval (λ,λ+∆λ).

The probability p(λ,R,r) will be affected by several factors which are included in the

equation5

( ) T( ) () ( ) ,R R,r

R

A

p ,R,r 2

o λ = ⋅ λ ⋅ ξ λ ⋅ ξ , (5.4)

where Ao/R2

 is the acceptance solid angle of the receiver optics with Ao being the area of

the telescope mirror, T(λ,R) is the atmospheric transmission factor at wavelength λ over

the range R, ξ(λ) is the spectral transmission factor of the receiver which includes the

effect of any spectral selecting elements such as filters and ξ(R,r) is the overlap factor or

the geometrical form factor defined as the probability of radiation from position r at the

range R being detected based on geometrical considerations. In this section, ξ(R,r) will be
89

assumed dependent only on the overlap area between the transmitted laser beam and the

receiver telescope field of view. In sections 5.1.1 and 5.1.2, this term will be discussed in

detail in terms of the receiver optics.

To evaluate the total power of the return signal collected by the receiver at the

instance t (where t = 2R/c represents the time interval between the propagation of the

laser pulse to the range R and the returned radiation to reach the receiver), equation (5.1)

must be integrated with respect to both the received wavelength and the range R.The

range integral is to account for the radiation reaching the receiver from any position along

the path of the laser pulse from which scattering occurs. The wavelength integral is to

consider the total receiver spectral window ∆λo centered at λ, which is usually defined by

the optical filter. Therefore, equation (5.1) can be expressed in the form5

( ) ( )( ) ( ) ∫ ∫ ∫ λ = λ λ ⋅ λ ⋅

∆λ

=

P , t dR d J ,R,r p ,R,r dA R,r

o

R ct 2

0

. (5.5)

Substituting equations (5.2) and (5.4) into equation (5.5) yields:

( ) () ( ) ( ) ( ) ( ) ( ) ∫ ∫ ∫ λ = ξ λ ⋅ λ β λ λ ⋅ λ ⋅ ξ ⋅ ⋅

∆λ

=

d , ,R,r T ,R R,r I R,r dA R,r

R

dR P , t A L

R ct 2

0

o 2

o

 (5.6)

The assumption that the observed radiation from the scattering medium is as narrow as

that of the laser radiation and that both are much smaller than the receiver spectral

window ∆λo, suggest that both Ii(λ) and β can be treated as delta functions. Furthermore,

assuming that the scattering medium is homogenous over the overlap between the field of

view and the laser beam results in equation (5.6) being written in the form:
90

( ) () ( ) ( ) ( )( ) ( ) ∫ ∫ λ = ξ λ β λ λ ⋅ λ ⋅ ξ ⋅ ⋅

=

R,r I R,r dA R,r

R

dR P , t A , ,R T ,R

R ct 2

0

o L 2 . (5.7)

As mentioned above, the assumption that the probability ξ(R,r) is unity where the field of

view of the receiver overlaps with the laser beam and zero elsewhere, and that the lateral

distribution of the laser pulse is uniform over an area AL(R) at the range R, results in

ξ( )( ) ( ) ()() () R,r ⋅ I R,r ⋅ dA R,r = ξ R ⋅I R ⋅ AL R ∫ . (5.8)

Equation (5.7) can be written as

( ) () ( ) ( ) ( ) ( ) ( ) ∫

=

λ = ξ λ β λ λ ⋅ λ ⋅ ξ ⋅ ⋅

R ct 2

0

o L L 2 R

dR P , t A , ,R T ,R R I R A R . (5.9)

For simplicity, assume that the temporal shape of the laser pulse is a rectangle of the

duration τL. Then, the limits of the range integration of equation (5.9) are c(t-τL)/2 to ct/2.

Furthermore, the range of interest is much larger than the pulse duration. Thus, the range

dependent parameters can be treated as constants over the small interval of the range

integration. Then, the total received power can be expressed by

( ) () ( ) ( ) ( ) ( ) ( ) 2

L

o L L R

c 2 P , t A , ,R T ,R R I R A R τ λ = ξ λ ⋅β λ λ ⋅ λ ⋅ ξ ⋅ ⋅ . (5.10)

For a rectangular laser pulse of duration τL, the intensity is given by

( ) ( )

A ( ) R

E T ,R I R

L L

L L

τ

λ = , (5.11)

where EL is the output energy of the laser pulse and T(λL,R) is the atmospheric

transmission factor at the laser wavelength to range R. From the Beer-Lambert law, the

transmission factors in equations (5.10) and (5.11) are given by
91

() () 













 λ = − κ λ ∫

R

0

L L T (5.12) ,R exp ,R dR

and

() () 













 λ = − κ λ ∫

R

0

T , (5.13) ,R exp ,R dR

where κ(λL,R) and κ(λ,R) are the atmospheric attenuation coefficients at the laser and

detected wavelengths, respectively. Combining equation (5.12) and (5.13) leads to the

total atmospheric transmission factor

() ( ) ( ) ( ) ( ) { } 













 = λ ⋅ λ = − κ λ + κ λ ∫

R

0

L L T . (5.14) R T ,R T ,R exp ,R ,R dR

The increment of radiation energy at wavelength λ received by the detector during

the interval (t,t+τd), where τd is the integration period for the detector, is given by

( ) () ∫

τ +

λ = λ

2R c

2R c

d

E . (5.15) ,R P ,t dt

combining equations (5.10), (5.11), (5.14) and (5.15) yields the scattered laser energy

received within the detector’s response time, τd:

( ) () ( ) ( ) ( ) 2

c , ,R

R

A E ,R E T R R d

2 L

o

L

τ λ = ξ λ ⋅ ⋅ ξ β λ λ . (5.16)

This is known as the basic scattering lidar equation.

For more general laser pulse shape, the average radiation power is given by

L

L

L

E P

τ = , (5.17)

and equation (5.16) will be modified to
92

( ) () ( ) ( ) ( ) 













 ⋅ − κ

τ λ = ξ λ ⋅ ξ ⋅β λ λ ∫

R

0

L

2 L

o

L exp R dR

2

c R , ,R

R

A P , (5.18) ,R P

where from equation (5.14)

κ() ( ) ( ) R = κ λ L ,R + κ λ,R . (5.19)

In the case of elastic (Mie or Rayleigh) scattering, the detected wavelength will be equal

to the laser wavelength and will be given by

( ) () ( ) ( ) ( ) 













 ⋅ − κ λ τ λ = ξ λ ⋅ ξ ⋅β λ ∫

R

0

L

L

2 L

o

L exp 2 ,R dR

2

c R ,R

R

A P . (5.20) ,R P

Equation (5.18) is the scattering lidar equation, while equation (5.20) is a special case

which is often used in the differential absorption lidar technique.

5.1.1. Receiver optics geometry

In the development of the scattering lidar equation the geometrical probability factor

ξ(R,r) was assumed unity wherever the field of view of the receiver optics overlaps the

laser beam and zero elsewhere. Also, the distribution of the laser intensity was assumed

uniform across the target plane. These assumptions are practical for long range

measurements. For short range measurements these factors must be considered.

A more general form of the laser intensity at the range R given in equation (5.11) is

( ) ( )

( ) ( ) ψ

π

λ

ψ = F R,r, W R

P T ,R I R,r, 2

L L , (5.21)

where r is the radial displacement of the point of interest in the target plane from the

telescope axis and ψ is the corresponding azimuth angle from a vertical plane passing

through this axis. As shown in Figure 5.1, W(R) is the radius of the laser pulse in the
93

target plane, and F(R,r,ψ) is the distribution of the laser power over the target plane at the

instant of interest. The Gaussian distribution is common in this case and therefore

( ) ( ) 





























 ψ = −

∗ 2

W R

r F , (5.22) R,r, exp

where

( )1 2 2 2 r = r + d − 2 ⋅r ⋅ d ⋅ cosψ ∗ (5.23)

and d represent the separation of the laser beam and telescope axis. Assuming a TEM00

mode laser, the radius of the laser beam is given by

( ) ( )1 2 2 2 2 W R = Wo + θ R , (5.24)

where Wo is the laser output aperture radius and θ is half of the laser divergence angle.

Also, the radius of the field of view of the telescope at range R is given by

rT ( ) R = ro + φR , (5.25)

where ro is the effective radius of the telescope mirror and φ is half of the field of view

angle of the telescope.

r (R) T

W(R)

R

d

Laser Beam Axis

Telescope Axis

Target Plane

Λ

Figure 5.1 Geometry of the target plane at range R.
94

Substituting equation (5.21) in equation (5.7) will yield the total scattered laser

power received by the detection system in the temporal duration τL and is given by

( ) () ( ) ()

( ) ( )( )

( )

∫ ∫∫

=

= = −τ

π

ψ=

× ξ ψ ⋅ ψ ⋅ ⋅ ψ β λ λ ⋅ λ = ξ λ

R ct 2

R c t 2

r

r 0

2

0

2 2

2 L

L o

L

T

dR R,r, F R,r, r drd

W R R

, ,R T R P , t P r .

 (5.26)

This equation can be further simplified considering that the range R is much greater than

the laser pulse length; therefore, the range dependent functions can be considered

constants in the small integration period. Thus, equation (5.26) can be written as

( ) ( ) ( ) ( )

( ) ( )( ) ∫ ∫

=

π

ψ=

ξ λ β λ λ × ξ ψ ⋅ ψ ⋅ ⋅ ψ

τ λ =

T

r

r 0

2

0

2 L 2

2

L o

L R,r, F R,r, r drd

W R

T R , ,R

R

r

2

c P . , t P

 (5.27)

In this case, the telescope effective area can be defined as

( ) ( ) ( )( ) ∫ ∫

=

π

ψ=

× ξ ψ ⋅ ψ ⋅ ⋅ ψ

π =

T

r

r 0

2

0

2

o R,r, F R,r, r drd

W R

A A , (5.28) R

where Ao = πro

2

. Therefore, the scattering lidar equation (5.18) can be re-written as

( ) ( ) () ( ) ( ) 













 ⋅ − κ

τ λ = ξ λ ⋅β λ λ ∫

R

0

L

L 2 L exp R dR

2

c , ,R

R

A R P (5.29) ,R P

in which A(R) in equation (5.29) has replaced Aoξ(R) in equation (5.18).

5.1.2. Geometrical form factor for simple overlap

Using equation (5.28), the definition of the geometrical form factor ξ(R) is given by

( ) ( ) ( )( ) ∫ ∫

=

π

ψ=

× ξ ψ ⋅ ψ ⋅ ⋅ ψ

π

ξ =

T

r

r 0

2

0

2 R,r, F R,r, r drd

W R

1 R . (5.30)
95

Assuming a uniform intensity distribution over the area of illumination at the range R and

considering a biaxial receiver system in which the laser axis is separated from the telescope

optical axis by a distance d, the geometrical form factor can be taken as a simple overlap

factor and can be written as

( ) { } () ()()

W ( ) R

r R ,W R ;d R R 2

T

π

Λ ξ = , (5.31)

where Λ is the area overlap function. The separation between the laser and telescope axes

in the target plane is

d = do − Rδ , (5.32)

where do is the separation at the lidar receiver and δ is the inclination angle between the

laser and the telescope axes.

The area overlap function assumes three cases as indicated in Figure 5.2. The first

(a) in which the telescope field of view does not overlap with the laser, the second (b) in

which there will be a partial overlap and the third (c) will be for full overlap. Therefore, Λ

will be given by5,84-85











π < −

ψ + ψ − ψ − < < +

> +

Λ =

r d r W

W r r d sin r W d r W

0 d r W

T

2

T

r T r T T

2

W T

2

T

, (5.33)

where















 + − ψ = −

2Wd

d W r

cos

2

T

2 2

1

W (5.34)

and















 + − ψ = −

2r d

d r W

cos

T

2 2

T

2

1

r . (5.35)
96

The situation of partial overlap is shown in Figure 5.3.

d

rT W

ψ r ψ W

d d

(a) (b) (c)

Figure 5.2 The three possible cases for telescope field of view and laser overlap functions,

(a) no overlap (b) partial overlap and (c) total overlap.

w

Telescope

Laser

do

R

θ

φ

δ

d

rT

Λ

r

o

wo

Figure 5.3 Geometry of the biaxial lidar illustrating the condition of partial overlap.

Introducing non-dimensional parameters as

or

R

z = , 

o

o

W

r A = , 

o

o

r

d D = , ρ( ) φ = = 1+ zφ

r

r

z,

o

T , ( ) δ = = D − zδ

r

d

s z,

o

,
97

( ) { }1 2 2 2 2

o

1 z A

W

W

π z,θ = = + θ , and ( ) ( )

( ) 2 2

2

z, A

z, y z, , ρ φ

π θ θ φ = ,

the overlap factor ξ(R) can be expressed by5,84-85

( ) ( )

( ) ( ) ( )

( ) ( )

 ( )













< −

− < < + 



 





ψ

ρ

ψ − π+

π

ψ

> +

ξ =

r R W d r W

sin z r W d r W

z

s z

z

y z

z 1

0 d r W

R

T

2 2

T

r r T T

W

T

, (5.36)

where

( ) () () () ()

()() () 















ρ

+ ρ − ρ ψ = −

2s z z y z

s z y z z z

z cos

2 2 2

1

W (5.37)

and

( ) () () () ()

()() 















ρ

+ ρ − ρ ψ = −

2s z z

s z z y z z

z cos

2 2 2

1

r . (5.38)

5.2. The DIAL Equation

Differential absorption lidar (DIAL) is a powerful technique to selectively measure

the density of molecules in the atmosphere as a function of altitude. It involves using two

laser pulses of slightly different wavelength. One is chosen to coincide with a strong

absorption feature of the specific molecular of interest and is known as the on-line

wavelength λon. The other is tuned to the side of the absorption feature where there is no

absorption and is known as the off-line wavelength λoff as illustrated in Figure 5.4.

The derivation of the DIAL measuring technique starts by considering the scattering

lidar equation (5.20) and modifying the total attenuation coefficient κ(λ,R)to

κ( ) ( ) ( )() λ,R ≡ κ λ,R + N R σ λ ' , (5.39)
98

where κ’(λ,R) is the attenuation coefficient exclusive of the absorption contribution from

the molecular of interest, N(R) is the number density of the molecular species at range R

and σ(λ) is the absorption cross section at wavelength λ. Therefore, applying the lidar

equation (5.20) at both the on and off-line wavelength respectively, we get

( ) ( )( )( ) [ ] ( ) () ( ) 















− κ λ + σ λ τ λ = ξ λ ξ β λ ∫

R

0

on on on

L

2 on on

o

on L exp 2 ,R N R dR

2

c R ,R

R

A P ,R P

 (5.40)

( ) ( )( )( ) [ ] ( ) () ( ) 















− κ λ + σ λ τ λ = ξ λ ξ β λ ∫

R

0

off off off

L

2 off off

o

off L exp 2 ,R N R dR

2

c R ,R

R

A P ,R P

 (5.41)

Wavelength

Water Vapor Absorption

Water Vapor 

Absorption Line OFF-LINE LASER

ON-LINE LASER

≈ 100 pm

Figure 5.4 DIAL on-line and off-line wavelengths relative to water vapor absorption line.

Then, the ratio of the return signals at the two wavelengths is obtained by dividing

equation (5.40) by (5.41) to obtain
99

( )

( ) ( ) 













 = − ∆σ

λ

λ

∫ exp 2 N R dR

P ,R

P ,R R

off 0

on , (5.42)

where the difference between the on and off-line wavelengths is less than 0.1 nm.

Therefore, one can assume that the receiver spectral transmission factor is independent of

wavelength over this small interval and similarly for the volume backscattering coefficients

and atmospheric attenuation coefficients. Also, the output laser power is assumed equal

for both wavelengths, and ∆σ = σon-σoff is the differential absorption cross section.

To evaluate the integration in equation (5.42), we use two successive ranges, R1 and

R2 respectively, to obtain

( )

( ) ( ) 













 = − ∆σ

λ

λ

∫ exp 2 N R dR

P ,R

P ,R R1

off 1 0

on 1 (5.43)

( )

( ) ( ) 













 = − ∆σ

λ

λ

∫ exp 2 N R dR

P ,R

P ,R R2

off 2 0

on 2 . (5.44)

Therefore,

( ) ( )( )

( )( )



 





λ λ

λ λ ∆σ = ∫

off 1 on 2

on 1 off 2

R

R P ,R P ,R

P ,R P ,R 2 N R dR ln

2

1

. (5.45)

The number density function can be assumed constant in the small interval ∆R=R2-R1

which defines the range cell. Hence, we get

( ) ( )( )

( )( )



 





λ λ

λ λ

∆σ∆ =

off 1 on 2

on 1 off 2

P ,R P ,R

P ,R P ,R ln

2 R

1 N R . (5.46)

Equation (5.46) is known as the differential absorption lidar or DIAL equation, and it is

used to obtain the number density of a specific molecular species as a function of range.
100

5.3. Applied Lidar Measurement Technique

In a typical lidar system, the return scattered radiation is collected by the receiver

telescope and then applied to a detector using focusing optics in order to convert the

optical signal into an electric one. Furthermore, the electric signal is digitized and stored in

digital form in a storage device. The stored data is available in the form of digitizer counts

versus number of samples, as shown in Figure 5.5. Knowing the digitization frequency fS

and considering the speed of light in air, the sample intervals can be transformed into

altitude or range using the range cell RC relation

S

C 2f

c R = . (5.47)

C(i)

C(i+1)

C(i+2)

S(i) S(i+1)

S(i+2)

Background

Samples

Counts

Roll-off point

Rise time effect

Figure 5.5 Ideal lidar return signal as obtained from the digitizer output.

The roll-off from the peak count in the return signal defines the beginning of the

useful data. The maximum altitude or range is obtained by considering the minimum

count, which corresponds to the receiver system signal-to-noise ratio, after subtracting the

signal background.
101

5.3.1. Aerosol measurement

To measure the aerosol density as a function of altitude, the visible and infrared laser

beams are used.86 These wavelengths are not absorbed but scattered by the molecular and

aerosols species; thus, we can use the backscattering lidar equation (5.20). The

atmospheric volume backscattering coefficient has the form of

( ) () () ( )

π

σ λ β λ ≈ β = 4 ,R R N R

s

L . (5.48)

To obtain the aerosol profile (relative measurements) equation (5.20) can be simplified to

() ( ) ( )

2 OT R

N R P R = constant ⋅ ⋅ , (5.49)

where the geometrical probability factor ξ(R) is considered unity for full overlap and OT is

the optical thickness defined by the exponential term of equation (5.20) considering the

atmospheric attenuation coefficients of scattering species87. Comparing the aerosol profile

to the standard atmospheric number density NSTD(R)88, results in the aerosol scattering

ratio expressed by

N() () () R = NSTD R ⋅ NSR R . (5.50)

Considering the digitizer gain Gd expressed in counts/(W/m2), the digitizer counts, Cd, can

be converted to the return signal power and the aerosol scattering ratio can be obtained by

( ) ( )

( )

O N ( ) R

C i R

G constant

1 N R

T STD

2

d

d

SR = ⋅ (5.51)

and

S

C

2f

R R = i ⋅ , (5.52)
102

where i is the sample number. To avoid the complexity of evaluating the constant term of

equation (5.51), the scattering ratio is usually normalized to the standard atmospheric

number density at a chosen altitude.

5.3.2. Water vapor measurement

Using the DIAL equation (5.46), the water vapor number density NWV(R) can be

obtained from the digitizer data according to

( ) ( ) ()

C () ( ) i C i 1

C i 1 C i

2 R

1 N R

off on

off on

WV ⋅ +

+ ⋅ ⋅ ∆σ∆ = (5.53)

and

S

C

2f

R R = i ⋅ , (5.54)

where Con and Coff are the on- and off-line return signal in counts. In order to smooth the

water vapor profile, averaging can be used for M data points where

( )

( )

M

C j

C i

M

j i '

∑= = . (5.55)

In this case, the range cell given in equation (5.47) will be modified to

M

2f

c R

S

C = . (5.56)
103

CHAPTER VI

VALIDATION OF THE DETECTION SYSTEM

The validation of the new water vapor detection system is discussed in this chapter.

The new detection system was connected to a telescope with focusing optics and filter to

construct a lidar receiver system. This receiver system was used to measure the

atmospheric return signals for relative aerosol and absolute water vapor density

measurements as a function of altitude. The aerosol measurement was obtained

simultaneously with NASA Langley Research Center ozone and aerosol DIAL system.6,84

The water vapor measurement was obtained by using a telescope and a laser transmitter

manufactured by Science and Engineering Services, Inc. (SESI).89-92 The measured water

vapor profile was compared with profiles obtained from Dulls Airport and Wallops Island

radiosondes.93

6.1. Receiver system setup

The new water vapor DIAL detection system was attached to a telescope and

focusing optics in order to build a lidar receiver system. The receiver is compatible with

radiation in the visible for aerosol measurement and near infrared region for water vapor

density measurements. A schematic diagram of this receiver system used for both aerosol

and water vapor measurements is shown in Figure 6.1.

Both 30.5 cm and 35.6 cm diameter telescopes were used to collect backscattered

radiation for aerosol and water vapor measurements respectively. These telescopes had a

Newtonian design with parabolic mirrors to focus the radiation and a flat mirror to turn
104

the focus on an optical fiber. A mechanism was used to allow the telescope to move in

two-dimension plane to focus the backscattered signal. The optical fiber had a 1 mm diasilica core and a doped silica clad of 1.1-mm with a 0.28 numerical aperture. This optical

fiber was custom designed by Cerme Optic, Inc. with an operating wavelength range of

200 to 1200 nm. The use of the optical fiber eliminates the need to align the focusing

optics with the telescope for compact receiver systems.

Focusing Optics Detection

System

Optical Fiber

Newtonian Telescope

L1 L2 F

Figure 6.1 Water vapor DIAL detection system as a part of the receiver.

The focusing optics consists of a plano-convex lens L1, a narrow-band interference

filter F and another plano-convex lens L2, as indicated in Figure 6.1. The filter was

changed according to the wavelength of the transmitted laser. The output beam is aligned

with the APD sensitive area using a three-dimension translation-stages mechanism to

move the detection system. Assuming a Gaussian beam distribution and considering the

diffraction effect, the light spot diameter at the focus dA containing 84% of the total

energy (Airy disk diameter) is defined by

D

F d 2.44 A

λ ⋅ = , (6.1)
105

where F and D are the focal length and the diameter of the focusing lens L2. The focal

spot diameter was checked for every operating conditions in order to insure that the APD

sensitive area is not over-flown by the incoming light.

The alignment of the focusing optics with the APD sensitive surface was tested by

simulating the lidar return using a stroboscope as a light source as shown in Figure 6.2.

The telescope collects the reflected light signal from the ceiling and the focusing optics

focuses the optical signal onto the APD. The trigger generator is used for synchronizing

the detection system with the stroboscope optical signal. This was done by triggering both

devices simultaneously with the delay unit used to adjust the collected signal in the

digitizer time window. The alignment was obtained by moving the detection system in

three-dimensions to give the optimum output signal from the detection system as shown in

Figure 6.3.

Power

Supply

Delay

Stroboscope

Clock

Trigger Computer

Telescope

Focusing Optics

Detection

System

Figure 6.2 Focusing optics alignment setup with the detection system.
106

50 100 150 200 250 300 350 400

2000

4000

6000

8000

10000

12000

14000

16000

Samples

Output in Counts

Figure 6.3 Detection system output corresponding to the best alignment.

6.2. Detection system validation with NASA airborne DIAL system

Simultaneous measurements of aerosol profiles were obtained using both the new

detection system and the NASA Langley Research Center airborne DIAL system, as

shown in Figure 6.4. The measurements were obtained during ground testing of the

system carried out on October 7, 1999.

Data Acquisition

System

Detector

System

Laser

Transmitter

Power

Supply

Delay

Clock

Computer

Focusing Optics

Detection

System

NASA Langley Research Center Airborne DIAL System

Precursor Signal

Figure 6.4 Detection system validation setup with NASA Langley Research Center

airborne DIAL system.
107

The NASA DIAL system consists of a laser transmitter, a receiver with a telescope

and a data acquisition system. The DIAL system measured aerosol backscattered profiles

at 598-, 622- and 1064-nm wavelengths. The visible wavelengths were compatible with

the new detection system and were used to validate it. The visible wavelengths were

separated by 300 μs with pulse energies of about 50 mJ for both wavelengths and a pulse

duration of 15 ns. The laser fire frequency was 30 Hz while the receiver digitization rate

was 1 MHz.

The new detection system was synchronized with the NASA DIAL system through

the laser precursor signal which defines the time duration between the two laser pulses.

This signal was applied to a delay stage, the output of which was used to trigger the new

detection system. This delay was applied in order to adjust the backscattered return signals

with the digitizer time window. The measured return signals from the new DIAL detection

system are shown in Figure 6.5a and b for the 589- and 622-nm wavelengths respectively.

The signals were obtained using 24 shot averages with 5-MHz digitization frequency

applied using a separate clock generator. The digitization frequency was reduced in order

to extend the time window to 1.64 ms. The 622-nm return signal shown in Figure 6.5

indicates an over-load which resulted in clipping of the return signal. As a result of the

clipping stage, this over-load did not affect the measurement. This was indicated in Figure

6.6a which shows the aerosol profiles obtained from both systems compared to the

atmospheric standard molecular density. The large aerosol return at lower altitude is due

to the atmosphere boundary layer. The disagreement between the measurements at lower

altitude may be due to the telescope misalignment in the near field region. Figure 6.6b

shows similar profiles obtained at the-598 nm wavelength. The aerosol profiles of the new
108

detection system where calculated by applying equations (5.51) and (5.52) using a 300 m

range cell.

0 200 400

0

2000

4000

6000

8000

10000

12000

14000

16000

Samples

CountsCounts

0 200 400

0

2000

4000

6000

8000

10000

12000

14000

16000

Samples

Clipping Action

(a) (b)

Figure 6.5 New detection system return signal obtained at (a) 598 nm and (b) 622 nm

with 24 shot average and 5 MHz digitization frequency.

Altitude in kmAltitude in km

0.5 1 1.5 2 2.5 3 3.5

0

2

4

6

8

10

12

Relative Aersole Number Density

New Detection 

System NASA

DIAL System

Standard Molecular Density

0.5 1 1.5 2 2.5 3 3.5

0

5

10

15

20

Relative Aerosol Number Density

New Detection 

System NASA

DIAL System

Standard Molecular Density

(a) (b)

Figure 6.6 Aerosol profiles obtained from the NASA DIAL system and

the new detection system at (a) 622 nm and (b) 598 nm

compared to the atmospheric standard molecular density.
109

6.3. Water vapor DIAL detection system validation

Science and Engineering Services, Inc. of Burtonsville MD, has developed a

compact diode-pumped Cr:LiSAF tunable laser to be used as the transmitter for water

vapor DIAL measurements from robotic aircraft. This system operates at 816-nm

wavelength with a maximum laser pulse energy of 25 mJ and a repetition rate of 5 Hz.

The new water vapor DIAL detection system was used with this transmitter. The

optical fiber cable and the focusing optics were used in conjunction with a 30 cm diameter

telescope to form a coaxial DIAL transmitter-receiver system. On November 5, 1999, this

DIAL system was used to measure the atmospheric water vapor profile. The return signals

for the on- and off-line are shown in Figure 6.7 for a 1000 shot average and 10-MHz

digitization frequency. The on- and off-line wavelengths are 815.3251 and 815.3278,

respectively, and the corresponding differential absorption cross-section is 5x10-24 cm

2

.

The laser energy was about 20 mJ per pulse. The water vapor profile was compared to

profiles obtained from radiosonde data from both Dulles Airport near Washington, D.C.,

and NASA Wallops, Wallops Island VA, as given in tables 6.1 and 6.2 respectively. Figure

6.8 shows the water vapor profiles from the radiosondes and the DIAL system obtained

using a 600 m range cell. The water vapor mass mixing ratio, r, obtained from the

radiosonde data where converted to number density, n, according to the relation

S

A r

M

N

n = ⋅ ⋅ρ , (6.2)

where NA is Avogadro’s number, M is the molecular weight of water which is equal to 18

kg/kmole and ρS is the air density. According to the ideal gas law, since the air density is a

function of both temperature and pressure, its value was obtained as a function of altitude
110

referring to the standard atmospheric data of the United States at zero altitude, where the

air pressure, temperature and density are 1.01325x103 mBar, 288.15 K and 1.225 kg/m3

respectively.87

200 400 600 800 1000 1200 1400 1600 1800

103

Altitude [1 sample = 30 m Alt.]

Counts

Counts vs Altitude

On-line signal

Off-line signal

Figure 6.7 On and off-line return signals.

-0.5 0 0.5 1 1.5 2 2.5 3 3.5

x 1017

0

200

400

600

800

1000

1200

1400

1600

1800

Water Vapor # density from DLS & WLPS sondes

Water Vapor Number Density in particle/cm^3

Altitude in m

Wallops Island radiosonde @ 12:00 pm

Dullas Airport radiosonde @ 12:00 pm

New DIAL Detection System @ 6:00 to 6:45 pm

Figure 6.8 Water vapor profiles obtained from the new system and radiosondes.
111

Table 6.1 Dulles Airport water vapor profile data on November 5, 1999 at 12:00 pm.

Altitude Temperature Pressure Density Water Vapor

M oC mBar kg/m3 gm/kg Cm-3

82 -2.3 1020 1.31193 2.89 1.3x1017

242 9.2 1000 1.23382 1.55 6.4x1016

309 13.8 992 1.20432 1.39 5.6x1016

893 9.2 925 1.14128 1.31 5.0x1016

1587 4.0 850 1.06842 1.00 3.6x1016

Table 6.2 Wallops Island water vapor profile data on November 5, 1999 at 12:00 pm.

Altitude Temperature Pressure Density Water Vapor

M oC mBar kg/m3 gm/kg cm-3

16 8.6 1030 1.2735 4.81 2.0x1017

80 9.8 1022 1.2483 4.58 1.9x1016

195 13.8 1008 1.2237 2.38 9.7x1016

262 13.4 1000 1.2157 2.33 9.5x1016

664 11.0 953 1.1684 1.47 5.7x1016

911 9.2 925 1.1413 1.20 4.6x1016

1606 4.0 850 1.0684 0.71 2.5x1016

Figure 6.8 indicates a good agreement between the DIAL system and the radiosonde

systems specially in the near-field. The deviation in the far-field might be due to the

difference in measurement location and time. Therefore, the expected return signal had to

be calculated using the lidar equation.

The measured return signals were compared to the expected return signal, calculated

from the lidar equation (5.16). Both signals are shown in Figure 6.9. To obtain this profile,

the geometrical form factor was considered for total overlap condition and was calculated

using equation (5.36). The transmitted laser beam had a 2-mm diameter with 10-3 radian

divergence angle and 3x10-3 radian telescope field of view. The total atmospheric

transmission factor was calculated considering the optical thickness and the standard

atmospheric number density given in equations (5.49) and (5.50) respectively. Good
112

agreement between the measured and calculated profiles, as shown in Figure 6.9, indicates

the validity of the new DIAL detection system.

0 500 1000 1500 2000 2

3

4

5

6

7

8

9

10

Altitude in m

Intensity in mW/m2

Off-line return signal

On-line return signal

Calculated return signal

Figure 6.9 Measured and calculated return signals.
113

CHAPTER VII

CONCLUSIONS

In this dissertation, an advanced water vapor DIAL detection system was designed,

constructed and evaluated. The system consists of an advanced APD detector, an analog

circuit and a digital circuit all on one, small size low mass, printed circuit board. The

system was interfaced to a personal computer. Both atmospheric aerosol and water vapor

concentration profiles were measured using this system.

The optical detector is a critical device in this system. A drift diffusion model was

developed for the APD to better understanding the device operation. This model is

suitable to solve for the device transient and steady-state operation assuming a four-layer

structure such as the reach-through and SLIK APD structures. An equivalent circuit was

also presented in order to model these devices in circuit simulation packages.

The APD choice was based on characterizing a group of such devices with different

structures from different manufactures. The characterization results indicate the better

performance of the SLIK structure over the reach-through structure in terms of noise and

device stability with respect to temperature. The selected advanced APD package had the

advantages of a built-in TEC and thermistor for temperature control and built-in transimpedance amplifier for direct current to voltage conversion.

The new detection system consists of an analog circuit and a digital circuit. The

analog circuit was designed using state-of-the-art electronics and controlled the APD in

terms of operating temperature and bias voltage. Also, it consists of a signal conditioning

stage, which adjusted the APD output to be compatible with the digitizer circuit.
114

The digital circuit consists of a digitizer, microcontroller and output circuit. The

digitizer was constructed using a state-of-the-art 14-bit, 10-MHz ADC, a dual port RAM

and a 13-bit counter to address the storage locations of the ADC output. The

microcontroller was used to synchronize the detection system with the laser fire pulse.

Also, it can perform real time averaging of the recorded data and monitor the APD

operating bias voltage and temperature. The output stage consists of a FIFO which is used

to collect the digital data and transfer it to a personal computer for storage.

The system design assumes an external power supply, clock and trigger signals. The

whole system was integrated onto one small printed circuit board suitable to be mounted

directly on the lidar receiver telescope. The final digital data is read using a parallel inputoutput interface with up to a 20 MHz data transfer rate. The operation of this interface is

completely controllable using software. The final data is stored in an ASCII file which is

accessible with any data handling software. The summary of the detection system

performance parameters is given in table 7.1.

The detection system validation started by reviewing the theory of the lidar remote

sensing technique with a special focus on the DIAL measuring technique. The

transformation from raw data representing the detection system output into a meaningful

atmospheric water vapor profile was also discussed. The integration of the detection

system into a complete receiver, using focusing optics and a telescope allowed the

validation of the system experimentally. One validation experiment was obtained by

measuring relative aerosol profiles simultaneously with the NASA Langley Research

Center airborne aerosol and ozone DIAL system. The other validation experiment was

obtained by integrating the new receiver system with a laser transmitter manufactured by
115

Science and Engineering Services Inc., and measuring water vapor profile which were

compared to Dulles Airport and Wallops Island water vapor radiosonde results. Both

experiments demonstrated the capability of the system in terms of range and measuring

atmospheric species other than water vapor.

Table 7.1 DIAL detection system performance summary.

End-to-end gain 183,969 counts/(W/m2

)

Maximum output 16383 counts

Minimum detectable signal @ 820 nm 0.85 μW/m2

Maximum detectable signal @ 820 nm 0.84 mW/m2

Dynamic range 1000

Maximum digitization frequency 10 MHz

Maximum time window 819.2 μs

Maximum repetition rate (8 k word) 10 Hz

Total Noise Equivalent Power (NEP) 212 fW/Hz1/2

Power consumption 21.1 W

Card size 15x9 cm2

Card weight 312 gm

Power supplies ± 5 V and ± 12 V

APD Sensitive area 0.238 mm2

Although the system performance is acceptable, future research is required for

additional improvements. One such improvement would be to replace the current ADC

with a 16-bit, 20-MHz ADC. The availability of fast computer interfaces suggests omitting

the microcontroller, which would result in even a more compact system. Also, more effort

is needed in developing the SLIK structure APDs with larger sensitive areas.

The good performance of the system suggested its application in aircraft based water

vapor DIAL systems such as the LASE instrument and space based DIAL systems such as

the visible channel of the Ozone Research with Advanced Cooperative Lidar Experiment

(ORACLE).
116

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125

APPENDIX A

APD MANUFACTURER DATA

AND CHARACTERIZATION RESULTS

A.1. APD designator and manufacturer data

APD Manufacturer Structure 1

Model Number d 2 R 3 VB 

4 In / Vn 5 BW 6

Serial Number T 7 λ 8 VBD 

9 Id 10 NEP 11

APD12 EG&G SLIK

C30649E 0.5 50 MV/W 349 150 nV/Hz1/2 11

147 25 820 356 7 0.003

APD11 EG&G SLIK

C30649E 0.5 50 MV/W 431 150 nV/Hz1/2 12

148 25 820 440 8 0.003

APD10 Electron Tubes, Inc. RTS

SSO-AD-2500 2.52 45 A/W 110 2.2x103

#8-12 21.5 880 161.7 30 0.4

APD9 Electron Tubes, Inc. RTS

SSO-ADH-500 0.5 45 A/W 150 1.3x103

#10 21.5 880 1.5 0.2

APD8 EG&G RTS

C30646E 0.5 20 MV/W 217 78 nV/Hz1/2 9.4 M

CD2507 22 850 228 0.004

APD7 EG&G RTS

C30646E 1.5 6 MV/W 384 120 nV/Hz1/2 9.5 M

CD2508 22 1064 443 0.02

 

1

 “SLIK” for super low κ structure, “RTS” for reach through structure and “BE” for beveled edge

structure.

2

 Diameter of the sensitive area in mm.

3

 APD responsivity.

4

 Bias Voltage in V.

5

 Spectral noise current and voltage density in nA/Hz1/2 and nV/Hz1/2 respectively. 6

 Bandwidth in MHz.

7

 Testing temperature in oC. 8

 Testing wavelength in nm.

9

 Breakdown voltage in V.

10 Dark current in nA.

11 Noise equivalent power in pW/Hz1/2.
126

APD6 EG&G RTS

C30659E 0.15 50 MV/W 420.4 110 nV/Hz1/2 15

CD2472 830 427 36 0.002

APD5 Advanced Photonix, Inc. BE

197-70-71-520 5 108 A/W 2 pA/Hz1/2 43

22 840 2450 160 0.014

APD4 EG&G RTS

C30950E 0.8 0.5 MV/W 275 15 nV/Hz1/2 50

25 830 425 0.029

APD2 EG&G RTS

C30955E 1.5 70 A/W 275 2 pA/Hz1/2 ---

1647 900 390 200 ---

A.2. SLIK APD package

RT

10k

TC 1STAGE

1020

C6

10nF

C7

10nF

C1

1000pF

0.5 mm

APD SLIK

CR1

HSNS-003

NE72000

B1

R2

150

R3

5.1K

R4

3.3K

R1

560k

R5

3.3K

R7

1K

R6

1K

R8

10

Q3

2N2857

Q4

2N4957

C4

10nF

C5

10nF

C3

10nF

C2

10nF;50V

Q2

2N4957

4

5

3

2

7

11

1

10

9

6

8

12

Thermal

Substrate

Detector Sensitive Area 0.24 mm2 (measured)

Trans-Impedance Amplifier Feedback Resistance 560 kΩ

Detector Maximum Power Loss 50 Mw

TEC Maximum Current 1.8 A

TEC Rated Current @ 0 0

C 0.5 A

TEC Area 3.96x3.96 mm2

Thermistor Resistance 



 



 



 





⋅ −

= ⋅ 298

1

T

1 3940

4

T R 10 e
127

A.3. APD calibration results

600 700 800 900 1000 1100

5

10

15

20

25

30

35

40

45

Wavelength in nm

Responsivity in A/W

Responsivity at

720 nm is 38.3 A/W

820 nm is 39.8 A/W

940 nm is 23.4 A/W

(a) APD12 Spectral Response at 336V and 25 C.

Responsivity at

720 nm is 45.6 A/W

820 nm is 50.1 A/W

940 nm is 30.7 A/W

600 700 800 900 1000 1100

5

10

15

20

25

30

35

40

45

50

55

Wavelength in nm

Responsivity in A/W

(b) APD11 Spectral Response at 415 V and 23 C.

(c) APD10 Spectral Response at 110V and 22.3 C.

600 700 800 900 1000 1100

0

5

10

15

20

25

30

35

40

45

Wavelength in nm

Responsivity in A/W

Responsivity at

720 nm is 37.1 A/W

820 nm is 39.8 A/W

940 nm is 21.3 A/W

(d) APD9 Spectral Response at 150V and 21.9 C.

Responsivity at

720 nm is 23.2 A/W

820 nm is 25.1 A/W

940 nm is 13.2 A/W

600 700 800 900 1000 1100

0

5

10

15

20

25

30

Wavelength in nm

Responsivity in A/W

600 700 800 900 1000 1100

0

1

2

3

4

5

6

7

8

9

Wavelength in nm

Responsivity in MV/W

Responsivity at

720 nm is 7.8 MV/W

820 nm is 8.4 MV/W

940 nm is 5.2 MV/W

(e) APD8 Spectral Response at 190V and 23 C. (f) APD7 Spectral Response at 380V and 23.6 C.

Fig.8 APD spectral response calibration.

Responsivity at

720 nm is 17.6 MV/W

820 nm is 23.5 MV/W

940 nm is 29.1 MV/W

600 700 800 900 1000 1100

5

10

15

20

25

30

Wavelength in nm

Responsivity in MV/W

o o

o o

o o
128

(g) APD6 Spectral Response at 410V and 22.2 C. (h) APD5 Spectral Response at 2350V.

(i) APD2 Spectral Response at 317V and 23.1 C.

600 700 800 900 1000 1100 20

30

40

50

60

70

80

90

Wavelength in nm

Responsivity in MV/W

Responsivity at

720 nm is 60.0 A/W

820 nm is 75.6 A/W

940 nm is 86.4 A/W

Fig.8 (continue) APDs spectral response calibration.

600 650 700 750 800 850 900 950 1000 15

20

25

30

35

40

45

Wavelength in nm

Responsivity in MV/W

Responsivity at

720 nm is 40.4 MV/W

820 nm is 40.3 MV/W

940 nm is 26.9 MV/W

Responsivity at

720 nm is 52.9 A/W

820 nm is 54.3 A/W

940 nm is 32.6 A/W

600 650 700 750 800 850 900 950 1000 10

15

20

25

30

35

40

45

50

55

60

Wavelength in nm

Responsivity in A/W

o

o

A.4. APD surface scan results

APD Active Area Area

mm

2 non-uniformity

APD11 0.2376 14.96 %

APD10 4.9550 18.76 %

APD9 0.1800 11.27 %

APD8 0.1820 16.62 %

APD7 1.5970 15.50 %

APD6 0.0148 17.64 %

APD5 18.88 7.40 %

APD4 0.5464 6.02 %

APD2 1.6920 6.10 %
129

A.5. APD responsivity variation with temperature

APD12 Bias Voltage : 336 V Condition : -3 < T< 25 oC

720 nm 2 3 3 116.3 8.69T 0.43T 8.3 10 T− − + − × A/W

820 nm 2 3 3 113.8 8.23T 0.41T 7.9 10 T− − + − × A/W

940 nm 2 3 3 57.2 3.65T 0.18T 3.6 10 T− − + − × A/W

APD11 Bias Voltage : 415 V Condition : -3 < T< 25 oC

720 nm 2 3 3 130.02 9.63T 0.45T 8.3 10 T− − + − × A/W

820 nm 2 3 3 136.19 9.85T 0.46T 8.6 10 T− − + − × A/W

940 nm 2 3 3 73.14 4.99T 0.24T 4.4 10 T− − + − × A/W

APD10 Bias Voltage : 110 V Condition : 6.3 < T< 22.3 oC

720 nm 3 2 2 3 4 3.85×10 −9.65×10 T + 90.53T −3.71T + 0.0557T A/W

820 nm 3 2 2 3 4 4.22×10 −10.62×10 T + 99.88T − 4.09T + 0.0617T A/W

940 nm 3 2 2 3 4 2.17×10 − 5.47×10 T + 51.55T − 2.12T + 0.0319T A/W

APD9 Bias Voltage : 150 V Condition : 10.4 < T< 21.9 oC

720 nm 2 2 3 3 58.9 2.92T 8.82 10 T 1.33 10 T − − − + × − × A/W

820 nm 2 2 3 3 66.8 3.53T 9.96 10 T 1.15 10 T − − − + × − × A/W

940 nm 2 2 4 3 35.2 1.96T 5.75 10 T 6.35 10 T − − − + × − × A/W

APD8 Bias Voltage : 190 V Condition : 5 < T< 23 oC

720 nm 7 5 3 2 1.5×10 − 4.9×10 T + 7.6×10 T V/W

820 nm 7 5 3 2 1.6×10 − 4.9×10 T + 7.58×10 T V/W

940 nm 7 5 3 2 8.95×10 − 2.6×10 T + 4.2×10 T V/W

APD7 Bias Voltage : 380 V Condition : 13.3 < T< 24.5 oC

720 nm 10 9 8 2 6 3 2.24×10 −3.26×10 T +1.63×10 T − 2.77×10 T V/W

820 nm 10 9 8 2 6 3 3.06×10 − 4.46×10 T + 2.24×10 T − 3.79×10 T V/W

940 nm 10 9 8 2 6 3 3.88×10 − 5.67×10 T + 2.86×10 T − 4.85×10 T V/W

APD6 Bias Voltage : 410 V Condition : 1.1 < T< 21.9 oC

720 nm 8 7 6 2 4 3 2.91×10 − 4.63×10 T + 3.07×10 T − 6.8×10 T V/W

820 nm 8 7 6 2 4 3 2.81×10 − 4.53×10 T + 3.04×10 T − 6.79×10 T V/W

940 nm 8 7 6 2 4 3 1.78×10 − 2.89×10 T +1.96×10 T − 4.4×10 T V/W

APD2 Bias Voltage : 317 V Condition : 5.7 < T< 23.1 oC

720 nm 2 3 4 928.5 −189.9T +17.1T − 0.6985T + 0.011T A/W

820 nm 2 3 4 1043.2 −192.3T +15.8T − 0.6005T + 0.0086T A/W

940 nm 2 3 4 1093.9 −185.9T +14.1T − 0.4956T + 0.0066T A/W
130

A.6. APD responsivity variation with bias voltage

APD12 Temperature : 25 oC Condition : 320 < V< 351 V

720 nm 7 5 3 2 3 3 4 1.88 10 2.25 10 V 1 10 V 2.02V 2 10 V− × − × + × − + × A/W

820 nm 7 5 3 2 3 3 4 1.89 10 2.27 10 V 1 10 V 2.04V 1.5 10 V− × − × + × − + × A/W

940 nm 7 5 2 2 3 4 4 1.01 10 1.22 10 V 5.47 10 V 1.09V 8.2 10 V− × − × + × − + × A/W

APD11 Temperature : 23 oC Condition : 390 < V< 430 V

720 nm 5 3 2 3 3 2.76 10 2.06 10 V 5.12V 4.25 10 V− − × + × − + × A/W

820 nm 5 3 2 3 3 2.91 10 2.17 10 V 5.41V 4.49 10 V− − × + × − + × A/W

940 nm 5 3 2 3 3 1.65 10 1.23 10 V 3.07V 2.54 10 V− − × + × − + × A/W

APD10 Temperature : 20.9 oC Condition : 80 < V< 115 V

720 nm 3 2 2 3 9.1 10 296.7V 3.19V 1.14 10 V− − × + − + × A/W

820 nm 3 2 2 3 9.5 10 308.6V 3.32V 1.19 10 V− − × + − + × A/W

940 nm 3 2 3 3 5.2 10 167.7V 1.81V 6.46 10 V− − × + − + × A/W

APD9 Temperature : 22 oC Condition : 125 < V< 150 V

720 nm 3 2 3 3 3.2 10 73.4V 0.55V 1.4 10 V− − × + − + × A/W

820 nm 3 2 3 3 3.6 10 80.5V 0.61V 1.5 10 V− − × + − + × A/W

940 nm 3 2 4 3 1.9 10 42.6V 0.32V 8 10 V− − × + − + × A/W

APD8 Temperature : 19.9 oC Condition : 100 < V< 215 V

720 nm 9 7 5 2 2 3 − 2.1×10 + 3.98×10 V − 2.5×10 V + 5×10 V V/W

820 nm 9 7 5 2 2 3 − 2.25×10 + 4.2×10 V − 2.6×10 V + 5.35×10 V V/W

940 nm 9 7 5 2 2 3 −1.4×10 + 2.69×10 V −1.66×10 V + 3.4×10 V V/W

APD7 Temperature : 22.1 oC Condition : 100 < V< 380 V

720 nm 7 6 3 2 3 2 4 3.81 10 9.19 10 V 7.83 10 V 27.1V 3.33 10 V− × − × + × − + × V/W

820 nm 7 6 4 2 3 2 4 5.07 10 1.22 10 V 1.04 10 V 36.1V 4.45 10 V− × − × + × − + × V/W

940 nm 7 6 4 2 3 2 4 6.28 10 1.52 10 V 1.29 10 V 44.7V 5.52 10 V− × − × + × − + × V/W

APD6 Temperature : 22.6 oC Condition : 100 < V< 350 V

720 nm 8 6 4 2 3 4 4.02×10 − 8.77 ×10 V + 6.80×10 V − 222.5V + 0.263V V/W

820 nm 8 6 4 2 3 4 4.61×10 −10.02×10 V + 7.75×10 V − 253.2V + 0.299V V/W

940 nm 8 6 4 2 3 4 2.96×10 − 6.35×10 V + 4.89×10 V −159.4V + 0.188V V/W

APD5 Temperature : 20 oC Condition : 2050 < V< 2350 V

720 nm 4 2 5 3 1.73 10 24.54V 0.012V 0.184 10 V− − × + − + × A/W

820 nm 4 2 5 3 1.75 10 24.79V 0.012V 0.186 10 V− − × + − + × A/W

940 nm 4 2 5 3 0.94 10 13.25V 0.006V 0.099 10 V− − × + − + × A/W

APD2 Temperature : 23.1 oC Condition : 310 < V< 340 V

720 nm 4 2 2 4 3 1.71 10 1.68 10 V 0.551V 6.05 10 V− − × + × − + × A/W

820 nm 4 2 2 3 3 3.28 10 3.16 10 V 1.018V 1.095 10 V− − × + × − + × A/W

940 nm 4 2 2 3 3 4.44 10 4.25 10 V 1.36V 1.45 10 V− − × + × − + × A/W
131

A.7. APD noise measurement

APD λ In NEP D*

nm fW/Hz1/2 cmHz1/2/W

720 1.9 2.87x1013

APD12 820 2.2x10-13 A/Hz1/2 1.9 3.05x1013

940 3.9 1.63x1013

720 1.8 2.77x1013

APD11 820 2.3x10-13 A/Hz1/2 1.7 2.90x1013

940 3.1 1.57x1013

720 74.3 2.99x1012

APD10 820 2.5x10-12 A/Hz1/2 69.3 3.21x1012

940 129.4 1.72x1012

720 80.7 5.26x1011

APD9 820 1.9x10-12 A/Hz1/2 74.9 5.67x1011

940 141.8 2.99x1011

720 11.8 3.62x1012

APD8 820 9.3x10-8 V/Hz1/2 11.0 3.88x012

940 17.7 2.41x1012

720 28.0 4.75x1012

APD2 820 1.7x10-12 A/Hz1/2 22.2 5.99x1012

940 19.4 6.84x1012
132

APPENDIX B

DETECTION SYSTEM APPLIED CIRCUITS

B.1. Analog-to-digital converter circuit

The AD9240 is a 10-M sample per second, 14-bit ADC. It has an on-chip low noise

sample and hold amplifier and a programmable voltage reference. Figure B1 shows the

associated programming circuit in our application. C32, C33, C34, C35 and Ca35 are 0.1-

μF bypass capacitors between each supply and its corresponding ground pin. The AD9240

maximum conversion rate can be controlled by R34. VINA and VINB are two analog

inputs for either a differential input mode or single-ended mode. By connecting VINB to

VREF and SENSE via R33, the second mode is selected with a common mode at 1 V

with 2 V input range. The chip internal reference, which appears on CAPT and CAPB, is

used in this connection. C28, C29, C30 and C31 form the recommended decoupling

network to insure a stable reference. It should be noted that this device provides the only

connection between the analog and the digital grounds.

SIGNAL

+5V +5VD

CLK

BIT 14

BIT 6

BIT 3

BIT 12

BIT 10

BIT 9

BIT 5

BIT 4

BIT 8

BIT 7

BIT 2

BIT 1

BIT 0

BIT 13

BIT 11

C32

0.1uf

Ra33

33

C29 +

10uf

C26

0.1uf

C27

0.1uf

U16

AD9240

7

8

9

10

11

12

13

14

15

16

17

18

19

20

21

22

23

24

25

26

27

30

29

2

1

5

33

31

32

42

39

36

37

41

34

38

40

43

44

35

3

28

4

6

CLK

N C

N C

N C

BIT 14

BIT 13

BIT 12

BIT 11

BIT 10

BIT 9

BIT 8

BIT 7

BIT 6

BIT 5

BIT 4

BIT 3

BIT 2

BIT 1

OTR

N C

N C

N C

AVSS

AVSS

DVSS

DRVSS

REFCOM

SENSE

VREF

VINB

CML

CAPB

CAPT

VINA

N C

N C

N C

N C

N C

BIAS

DVDD

AVDD

AVDD

DRVDD

C30

0.1uf

C31

0.1uf

R34

2k

R33

33

C35

0.1uf

Ca35

0.1uf

C34

0.1uf

U20A

74LS04

1 2

C33

0.1uf

+

C25

10uf

C28

0.1uf

Figure B1 Analog to digital converter circuit.

B.2. Data collect and store circuit

The data collect and store circuit acts as a single shot transient recorder. The

operation of this circuit is fully controlled by the microcontroller through the “counter

enable” signal, applied by the two input NAND gate U15A. A microcontroller “clear

counter” signal clears the contents of all three, four bit binary counters and the J-K flip

flop. The maximum count is detected by four input NAND gate, U14A, which
133

automatically stops the counting operation through U15A and sends an “end-of-count”

signal to the microcontroller.

To reduce pick-up noise, the inverter U20A enables the data conversion by the ADC

in a different half cycle than that of the counter. Connecting the MSB of the RAM left

address to supply enable the data to be stored only in the lower half of the memory.

CLK

Counter Enable

Clear Counter

SIGNAL

+5VD

CLK

+5VD

End of Count

CLK

BIT 5

AL1

AL7

Q11

Q 7

Q12

BIT 6

BIT 7

BIT 3

Q 9

Q13

AL6

BIT 0

Q 8

AL11

Q11

BIT 2

AL9

AL5

BIT 1

Q 9

BIT 4

Q14

AL0 AL11

AL7

Q 8

BIT 2

BIT 3

AL3

AL10

Q14

AL9

AL8

AL1

AL2

AL4

Q12

AL2

BIT 1

Q 7

Q10

BIT 7

BIT 5

BIT 4

AL4

BIT 6

AL0

Q13

AL3

AL10

BIT 0

AL5

AL6

AL8

Q10

BIT 10

Q 4

BIT 13

Q 3

Q 2

Q 1

BIT 14

Q 5

Q 5

BIT 12

BIT 12

Q 6

Q 3

Q 2

Q 1

BIT 9

BIT 14

BIT 13

BIT 11

Q 0

BIT 8

Q 6

Q 0

BIT 11

BIT 8

BIT 10

Q 4

BIT 9

U11

74191

1 5

1

1 0

9

1 4

4

5

1 1

3

2

6

7

1 3

1 2

A

B

C

D

CLK

G

D/U

LOAD

Q A

Q B

Q C

Q D

RCO

MX/MN

U12

74191

1 5

1

1 0

9

1 4

4

5

1 1

3

2

6

7

1 3

1 2

A

B

C

D

CLK

G

D/U

LOAD

Q A

Q B

Q C

Q D

RCO

MX/MN

U15A

12

7400

3

U13A

74112

4

1

1 6

1 5

1 4

2 3

J

CLK

K

Q

Q

P R C L

U10

74191

1 5

1

1 0

9

1 4

4

5

1 1

3

2

6

7

1 3

1 2

A

B

C

D

CLK

G

D/U

LOAD

Q A

Q B

Q C

Q D

RCO

MX/MN

U17

74541

2

3

4

5

6

7

8

9

1

1 9

1 8

1 7

1 6

1 5

1 4

1 3

1 2

1 1

A1

A2

A3

A4

A5

A6

A7

A8

G 1

G 2

Y 1

Y 2

Y 3

Y 4

Y 5

Y 6

Y 7

Y 8

U20A

7404

1 2

U19

IDT7026

1

2 1

8 3

2

6 6

6 7

6 8

6 9

7 0

7 1

7 2

7 3

7 4

7 5

7 6

7 7

7 8

7 9

8 0

3

4

6

7

8

9

1 0

1 1

1 2

1 3

1 4

1 5

1 6

1 7

1 9

2 0

8 4

6 5

8 1

8 2

2 2

1 8

5

Vcc

Vcc

SEML

OEL

A0L

A1L

A2L

A3L

A4L

A5L

A6L

A7L

A8L

A9L

A10L

A11L

A12L

A13L

LBL

I/O0L

I/O1L

I/O2L

I/O3L

I/O4L

I/O5L

I/O6L

I/O7L

I/O8L

I/O9L

I/O10L

I/O11L

I/O12L

I/O13L

I/O14L

I/O15L

R/WL

BUSYL

UBL

CEL

GND

GND

GND Ra33

3 3

U16

AD9240

7

8

9

1 0

1 1

1 2

1 3

1 4

1 5

1 6

1 7

1 8

1 9

2 0

2 1

2 2

2 3

2 4

2 5

2 6

2 7

3 0

4 1

CLK

N C

N C

N C

BIT 14

BIT 13

BIT 12

BIT 11

BIT 10

BIT 9

BIT 8

BIT 7

BIT 6

BIT 5

BIT 4

BIT 3

BIT 2

BIT 1

OTR

N C

N C

N C

VINA

U14A

7420

1 2

4

5

6

U18

74541

2

3

4

5

6

7

8

9

1

1 9

1 8

1 7

1 6

1 5

1 4

1 3

1 2

1 1

A1

A2

A3

A4

A5

A6

A7

A8

G 1

G 2

Y 1

Y 2

Y 3

Y 4

Y 5

Y 6

Y 7

Y 8

Figure B2 Data collect and store circuit.

B.3. Microcontroller connection circuit

The 80C196KB-16 microcontroller shares the data and address on one 16 bit bus.

The address latch, formed by U24 and U25, is used to distinguish and separate the address

and the data words. The inverter U20B is used to divide the microcontroller memory into

two sections. One section accesses the right address of the dual port RAM, while the

other section is sub-divided into two sub-sections. The two sub-sections are decoded to

access the FIFO using the four input AND gate U26A, while the other is used to access

the 8-bit dip-switch using the four input AND gate U26B and the inverters U20C and

U20D. The control of the FIFO and counter operations is obtained using the

microcontroller port 1. Port 2 is used to sense the “trigger” signal and for serial

communication if applied. Port 0 is set as analog channels to the internal analog to digital
134

converter in order to read the APD temperature and bias voltage values. Each monitor

channel is protected using the interface circuit shown in figure B4a. Figure B4b shows the

microcontroller reset circuit.

RESET

TRIG

+5V

FIFO Write

+5VD

+5VD

+5VD +5VD

FIFO Enable

Read

FIFO Data

Temp. Monitor

Bias Monitor

FIFO Ready

FIFO Empty (a)

FIFO Empty (b)

End of Count

FIFO Reset

Counter Enable

Clear Counter

Serial Out

Serial In

Read

AR5

AR6

AR6

AR5

D14

D12

D12

AR10

D11

AR9 D10

D11

D14

D15

D12

D14

D13

D13

AR7

D15

D11

D13

AR8

D15

AR10

AR8

AR7

D10 D10

AR9

D9

D9

D8

D8 D8

D9

AR12

AR12

AR11

AR13

AR11

AR13

AR12

AR11AR13

W 1

W 2

W 6

W 8

W 3

D4

W 3

W 2

D6

D1

D2

D5

D0

W 4

W 1

W 3

D0

D6

D3

D1

D4

D2

W 6

W 5

W 6

W 5

W 7

D4

W 8

D3

D4

W 8

D5

W 1

D7

D7

D0

W 4

D7

D1

D3

W7 W7

W 4

D6

D2

W 5

D5

D5

W 2

D7

D3

D6

D2

D1

AR0

AR0

AR4

AR4

AR3

AR3

AR2

AR1

AR1

AR2

U15C

74LS00

9

10

8

U20E

74LS04

11 10

U26A

74LS21

1 4

2

5

6

U26B

74LS21

9

1210

13

8

U22

74ALS573

2

3

4

5

6

7

8

9

11

1

19

18

17

16

15

14

13

12

D1

D2

D3

D4

D5

D6

D7

D8

C

OC

Q1

Q2

Q3

Q4

Q5

Q6

Q7

Q8

U20C

74LS04

6 5

U20B

74LS04

3 4

U21

74ALS573

2

3

4

5

6

7

8

9

11

1

19

18

17

16

15

14

13

12

D1

D2

D3

D4

D5

D6

D7

D8

C

OC

Q1

Q2

Q3

Q4

Q5

Q6

Q7

Q8

U20D

74LS04

9 8

Y1

20 MHz

+ C36

20 pf

C37 +

20 pf

U23

80C196KB

60

59

58

57

56

55

54

53

52

51

50

49

48

47

46

45

1

13

14

68

36

12

37

67

66

16

65

6

5

7

4

11

10

8

9

19

20

21

22

23

30

31

32

18

17

15

44

42

39

33

38

40

41

61

2

3

62

24

25

26

27

28

29

34

35

64

43

63

AD0

AD1

AD2

AD3

AD4

AD5

AD6

AD7

AD8

AD9

AD10

AD11

AD12

AD13

AD14

AD15

Vcc

Vref

Vss

Vss

Vss

ANGND

Vpp

XTL1

XTL2

RESET

CLKOUT

P0.0

P0.1

P0.2

P0.3

P0.4

P0.5

P0.6

P0.7

P1.0

P1.1

P1.2

P1.3

P1.4

P1.5

P1.6

P1.7

TXD/P2.0

RXD/P2.1

P2.2

P2.3

P2.4

P2.5

P2.6

P2.7

W R

BHE

RD

EA

NMI

ALE

SID0

SID1

SID2

SID3

HSO0

HSO1

HSO2

HSO3

BUSWIDTH

READY

INST

U24

74ALS573

2

3

4

5

6

7

8

9

11

1

19

18

17

16

15

14

13

12

D1

D2

D3

D4

D5

D6

D7

D8

C

OC

Q1

Q2

Q3

Q4

Q5

Q6

Q7

Q8

S1

SW DIP-8

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

U25

1.1k

1 16

2 15

3 14

4 13

5 12

6 11

7 10

8 9

U19

IDT7026

26

63

44

41

42

23

24

25

27

28

29

30

31

32

33

34

35

36

37

38

40

45

39

43

64

46

47

62

48

49

50

51

52

53

54

55

56

57

58

60

61

59

Vcc

M/S

SEMR

OER

R/WR

I/O0R

I/O1R

I/O2R

I/O3R

I/O4R

I/O5R

I/O6R

I/O7R

I/O8R

I/O9R

I/O10R

I/O11R

I/O12R

I/O13R

I/O14R

I/O15R

CER

GND

GND

GND

UBR

LBR

BUSYR

A13R

A12R

A11R

A10R

A9R

A8R

A7R

A6R

A5R

A4R

A3R

A1R

A0R

A2R

Figure B3 Microcontroller connection circuit.
135

+5VD

RESET

D 3

1N4148

S2

RESET

R37

33k

C38

22uF

+5V

MONITOR IN MONITOR OUT

R44

270

C39

0.005uF

D 5

1N4148

D 4

1N4148

(a) (b)

Figure B4 Microcontroller (a) ADC interface and (b) reset circuit.

B.4. Output stage

The FIFO output data is buffered using U29 and U30, the output of which is applied

to the four channel opto-couplers U31-U34 for ground optical isolation. The FIFO control

signals “FIFO ready” and “read FIFO” are also optically isolated using the dual separate

channels U37 optocouplers.

+5VD

+5VD +5VD

+5VD

+5VD

+5VD

FIFO Data

FIFO Ready

FIFO Reset

FIFO Enable

FIFO Write

FIFO Empty (a)

FIFO Empty (b)

+5VD

Serial In

Serial Out

B6 B6

B13

H7

B4

D4

D3

D2

DATA0

DATA0

DATA14

R13

H1

B2

B9

H2

H10 H9

B13

D1

R12

DATA11

R14

H5

H13

B0

DATA7

DATA2

DATA10

B10

H0

H12

B7

D14

DATA11

R10

DATA12

B5

B9

H13

H8

B12

H11

DATA1

R13

DATA6

B10

H7

B14

D5

R4

R15

R8

DATA14

DATA7

H4

B8

D9

DATA6

DATA10

R1

R6

R0

B0

H9

B8

H2

DATA3

DATA9

R0

R8

B4

H1

H8

H3

D11

D0

R3

R9

DATA5

R1

R2

R5

H15

H4

B1

D10

R2

R11

B15

B15

B14

B3 B3

DATA8

DATA3

DATA8

R15

H14

H12

B11

D15

D8

DATA5

DATA1

DATA4

R10

DATA12

R7

H6

B12

D12

DATA4

R14

R9

DATA15

DATA13

H14

B2

B5

D13

R5

DATA9

R3

B11

H11

H15

D6

R6

R12

H6

H3

B7

R4

B1

H0

H5

H10

D7

DATA2

R7

R11

DATA13

DATA15

U14B

74LS20

9

1012

13

8

U36

250

1 16

2 15

3 14

4 13

5 12

6 11

7 10

8 9

U33

HCPL-6650

2

3

7

6

10

13

14

15

12

11

9

16

1

4

5

8

I1

I2

I4

I3

GND

OP2

OP1

Vcc

OP3

OP4

NC

NC

O1

O2

O3

O4

U34

HCPL-6650

2

3

7

6

10

13

14

15

12

11

9

16

1

4

5

8

I1

I2

I4

I3

GND

OP2

OP1

Vcc

OP3

OP4

NC

NC

O1

O2

O3

O4

U32

HCPL-6650

2

3

7

6

10

13

14

15

12

11

9

16

1

4

5

8

I1

I2

I4

I3

GND

OP2

OP1

Vcc

OP3

OP4

NC

NC

O1

O2

O3

O4

U31

HCPL-6650

2

3

7

6

10

13

14

15

12

11

9

16

1

4

5

8

I1

I2

I4

I3

GND

OP2

OP1

Vcc

OP3

OP4

NC

NC

O1

O2

O3

O4

U35

250

1 16

2 15

3 14

4 13

5 12

6 11

7 10

8 9

U15D

74LS00

12

13

11

U30

74LS541

2

3

4

5

6

7

8

9

1

19

18

17

16

15

14

13

12

11

A1

A2

A3

A4

A5

A6

A7

A8

G1

G2

Y1

Y2

Y3

Y4

Y5

Y6

Y7

Y8

U28

IDT7205

6

5

4

3

27

26

25

24

2 28

9

10

11

12

16

17

18

19

13

14

7

8

1

15

22

21

23

20

D0

D1

D2

D3

D4

D5

D6

D7

D8 Vcc

Q0

Q1

Q2

Q3

Q4

Q5

Q6

Q7

Q8

GND

XI

FF

W

R

RS

EF

FL/RT

XO/HF

U27

IDT7205

6

5

4

3

27

26

25

24

2 28

9

10

11

12

16

17

18

19

13

14

7

8

1

15

22

21

23

20

D0

D1

D2

D3

D4

D5

D6

D7

D8 Vcc

Q0

Q1

Q2

Q3

Q4

Q5

Q6

Q7

Q8

GND

XI

FF

W

R

RS

EF

FL/RT

XO/HF

U29

74LS541

2

3

4

5

6

7

8

9

1

19

18

17

16

15

14

13

12

11

A1

A2

A3

A4

A5

A6

A7

A8

G1

G2

Y1

Y2

Y3

Y4

Y5

Y6

Y7

Y8

J1

68 SCSI

10

44

45

12

13

47

48

15

16

17

51

52

53

54

21

22

1

3

2

11

PA0

PA1

PA2

PA3

PA4

PA5

PA6

PA7

PB0

PB1

PB2

PB3

PB4

PB5

PB6

PB7

Vcc

READY(ACK1)

READ(REQ1)

GND

U15B

74LS00

45

6

R38

TBD U20F

74LS04

12 13

R39

TBD

R36

250

R35

250

U37

HCPL-6630

19

20

8

7

10

15

13

12

3

2

1

4

5

6

9 11

14

16

17

18

+

-

Vo1

GND1

Vcc1

Vcc2

Vo2

GND2

+

-

NC

NC

NC

NC

NC NC

NC

NC

NC

NC

J2

9 Pin Connector

5

9

4

8

3

7

2

6

1

Figure B5 Microcontroller connection circuit.
136

B.5. Chip power supply bypassing

Generally, bypass or coupling capacitors are used with every supply pin of any

device. It is used to supply the device with inrush current required at switching instants.

Figure B6 shows the used power supply bypassing for an op amp, for both the positive

and the negative supplies. Inductors Lx1 and Lx2 are used for smoothing the device bias

current. As a layout restriction, ceramic capacitors (Cx3 and Cx1 are 0.1 μF) must be

located less than 0.3 cm away from the supply pin. Cx2 and Cx4 are 10 μF electrolyte

capacitors. For all devices other than op amps, 0.1 μF capacitors are only in use to couple

every supply pin to the ground.

+5V

-5V

OUTPUT

INPUT 1

INPUT 2

Cx1

0.1uF

Lx2

BLM41A151S

Cx3

0.1uF

+ Cx2

10uF

+

Cx4

10uF

+

-

Uxx

Op Amp

3

2

6

7 4

Lx1

BLM41A151S

Figure B6 Op amp connection to power supplies.

B.6. Clock and trigger isolation

U38, the HCPL-2430, is a dual channel, TTL compatible optocoupler. It is used to

optically isolate the clock and the trigger inputs as shown in Fig.D5. Resistors R41 and

R42 are 50Ω termination resistors. R40 and R43 are used to limit light emitting diode

current.

+5VD

CLK

TRIG

R40

625

R43

625

R41

50

R42

50

U38

HCPL-2430

1

2

3

4

5

6

7

8 +

-

-

+

GND

OP2

OP1

Vcc CLOCK2

TRIGGER1

TRIGGER2

CLOCK1

Figure B7 External clock and trigger isolation circuit.
137

APPENDIX C

ELEMENTS OF THE NEW DETECTION SYSTEM

C.1. Detection system components

Referring to Chapter 4 and Appendix B, this appendix gives the component list of

the new detection system along with the description of these component, the manufacturer

and the number of parts used.

Component Part Reference Description Manufacturer # of

Parts

AD586BR U1 Reference Voltage Analog Devices 1

AD706AR U2,U3,U5 Dual Op-Amp Analog Devices 3

521-5-M U4 High Voltage Supply Analog Modules 1

C30649E U6 Avalanche Photodiode EG&G 1

AD8036AR U7 Clipping Amplifier Analog Devices 1

AD8041AR U8 Buffer Amplifier Analog Devices 1

B3LL31-25-P/P U9 Output Filter Microwave Inc. 1

74F191SC U10,U11,U12 4 Bit Counter FAIRCHILD 3

TC74AC112FN U13 Dual JK Flip Flop TOSHIBA 1

TC74VHC20FT U14 Dual, 4 input NAND TOSHIBA 1

TC74VHC21FT U26 Dual, 4 Input AND TOSHIBA 1

TC74VHCT00AFT U15 Quad,2 Input NAND TOSHIBA 1

AD9240AS U16 14 Bit, 10 MHz ADC Analog Devices 1

TC74VHCT541AF

T

U17,U18,U29,U30 Octal Buffers TOSHIBA 4

IDT7026L20J U19 Dual Port RAM IDT 1

TC74VHC04FT U20 Hex Inverters TOSHIBA 1

TC74VHC573FT U21,U22,U24 8 Bit, Address Latch TOSHIBA 3

N87C196KB-16 U23 Microcontroller INTEL 1

IDT7205L15J U27,U28 8k, 9Bit, FIFO IDT 2

HCPL-6650 U31,U32,U33,U34 4 Channel Optocoupler Hewlett Packard 4

HCPL-6630 U37 2 Channel Optocoupler Hewlett Packard 1

HCPL-2430#300 U38 2 Channel Optocoupler Hewlett Packard 1

766-163-R(1k) U25 1k, 2%,160mW,

Resistor Network

CTS 1

766-163-R(270) U35,U36 270, 2%,160mW,

Resistor Network

CTS 2

LL4148DICT D1,D3,D4,D5,D6,

D7

Diode, 1N4148 VISHAY 6

BZX84C5V1DICT D2 5.1V, Zener Diode VISHAY 1

TIP110 Q1 Darlington Transistor Texas Instrument 1

NFM46P11C155 L1,L2,L4,L5,L16 Line Filter MURATA 5

BLM41A151S L3,L6,L7,L8,L9,

L10,L11,L12,L13,

L14,L15

Chip Ferrite Bead

Inductors

MURATA 11
138

Component Part Reference Description Manufacturer # of

Parts

1812Y104JXE C8 0.1μF, 500 V, 5%

Ceramic Capacitor

VISHAY 1

VJ1812Y105JXXM C2,C10,C5,C5a 1μF, 25V, 5%

Ceramic Capacitor

VISHAY 4

PCC472BCT C39,C40 5nF, 50V, 10%,

Ceramic Capacitor

Panasonic 2

PCS1226CT C38 22uF, 6.3V

Tantalum Capacitor

Panasonic 1

PCC200CCT C36,C37 20pF, 50V, 5%

Ceramic Capacitor

Panasonic 2

PCC220JCT C9 22pF, 3kV, 5%

Ceramic Capacitor

Panasonic 1

PCS3106CT C14,C17,C21,C23,

C25, C29,C3u23

10μF, 16V

Tantalum Capacitor

Panasonic 6

PCC103BCT Cu31,Cu32,Cu33,

CU34,C1u37,C2u37

C1u19,C2u19,

C3u19,C1u23

0.01μF, 50V, 10%

Coupling Capacitor

Panasonic 6

PCC104BCT C1,C3,C4,C6,C7,

C11,C12,C13,C15,

C16,C18,C19,C20,

C22,C24,C26,C27,

C28,C30,C31,C32,

C33,C34,C35,Cu10,

Ca35,Cu11,Cu12,

Cu13,Cu14,Cu15,

Cu17,Cu18,Cu20,

Cu21,Cu22,Cu24,

Cu26,Cu27,Cu28,

0.1μF, 50V, 10%

Coupling Capacitor

Panasonic 42

895-0205 R32 5 Ω, 1% Resistor Dale 2

P33ECT R33,Ra33 33Ω, 5% Resistor Panasonic 2

P50FCT R21,R30,R31,R41,

R42

49.9 Ω, 1% Resistor Panasonic 5

TNPW1206100B R26 100 Ω, 1% Resistor VISHAY 2

P130FCT R22 130 Ω, 1% Resistor Panasonic 1

P140FCT R23 140 Ω, 1% Resistor Panasonic 1

P270ECT R35,R36,R44,R45 270 Ω, 5% Resistor Panasonic 4

TNPW1206301B R27,R28,R29 301 Ω, 1% Resistor VISHAY 3

P499FCT R6 499 Ω, 1% Resistor Panasonic 1

P620ECT R40,R43 620Ω, 5% Resistor Panasonic 2

P2KZCT R14,R34 2 kΩ, 0.1% Resistor Panasonic 2

P3KZCT R2 3 kΩ, 0.1% Resistor Panasonic 1

P10KZCT R1,R8,R24,R25 10 kΩ, 0.1% Resistor Panasonic 4

P22KZCT R13 22 kΩ, 0.1% Resistor Panasonic 1

P33KZCT R17,R15,R16,R37 33 kΩ, 0.1% Resistor Panasonic 4

P49.9KFCT R3 49.9 kΩ, 1% Resistor Panasonic 1

P82KZCT R12 82 kΩ, 0.1% Resistor Panasonic 1
139

Component Part Reference Description Manufacturer # of

Parts

P200KFCT R4,R5,R19 200 kΩ, 1% Resistor Panasonic 3

R9,R10,R11 2.55 MΩ Resistor 3

R18,R38,R39 TBD

SM4W103 R7,R20 10 kΩ, POT, 10% Philips 2

SPE1211 JP1,JP2,JP3,JP4 Jumpers Specialty Elec.Inc. 2

777600-01 J1 68 Pin Connector AMP 1

AFR09G J2 9 Pin D Connector AMP 1

AMR09G J3 9 Pin D Connector AMP 1

CT2198MST S1 8 Pin, Dip Switches CTS 1

P8087SCT S2 Push Button Switch Panasonic 1

XC550CT Y1 16 MHz Crystal ECS 1

SK-PLCC68-SO1 68pin, PLCC Socket Ironwood Elec. 1

C.2. Detection system power consumption

The steady state maximum power consumption of the individual components of the

detection system and the total power for each section are given as follows:

Reference Voltage

P U1 500.00 mW

P U2 650.00 mW

P R1 2.50 mW

P R2 0.75 mW

Total power P1 1153.25 mW (1)

Signal Conditioning

P U6 250.00 mW

P U7 900.00 mW

P U8 900.00 mW

P R21 45.00 mW

P R24 0.70 mW

P R25 0.70 mW

P R26 22.50 mW

P R27 67.50 mW

P R28 67.50 mW

P R29 67.50 mW

P R30 101.30 mW

P R31 101.30 mW

Total power P2 2524.00 mW (2)
140

Voltage Controller

P U3 650.00 mW

P U4 480.00 mW

P R3 0.50 mW

P R4 2.00 mW

P R5 2.00 mW

P R6 50.00 mW

P R7 2.50 mW

P R8 0.10 mW

P R9-11 20.70 mW

P R12 0.20 mW

P R13 0.10 mW

Total power P3 1208.10 mW (3)

Temperature Controller

P U5 650.00 mW

P R14 24.50 mW

P D2 16.70 mW

P R15 0.40 mW

P R16 0.40 mW

P R17 0.40 mW

P R19 0.20 mW

P R20 14.40 mW

P RT 0.40 mW

Total power P4 707.40 mW (4)

P R32 625.00 mW

P Q1 1875.00 mW

P TEC 125.00 mW

Total power P5 2625.00 mW (5)

Data Collect and Store

P U10 100.00 mW

P U11 100.00 mW

P U12 100.00 mW

P U13 180.00 mW

P U14 180.00 mW

P U15 180.00 mW

P U16 285.00 mW

P U17 180.00 mW

P U18 180.00 mW

Total power P6 1485.00 mW (6)
141

Microcontroller Circuit

P U19 750.00 mW

P U20 180.00 mW

P U21 180.00 mW

P U22 180.00 mW

P U23 1500.00 mW

P U24 180.00 mW

P U25 181.80 mW

P U26 180.00 mW

Total power P7 3331.80 mW (7)

Output Stage

P U27 770.00 mW

P U28 770.00 mW

P U29 180.00 mW

P U30 180.00 mW

P U31 800.00 mW

P U32 800.00 mW

P U33 800.00 mW

P U34 800.00 mW

P U35 800.00 mW

P U36 800.00 mW

P U37 400.00 mW

P U38 700.00 mW

P R40 10.00 mW

P R41 125.00 mW

P R42 125.00 mW

P R43 10.00 mW

Total power P8 8070.00 mW (8)

Total Power for Analog Circuit from ±5 V Supply

=P2+P5= 2524.00+2625.00 =5149.00 mW

Total Power for Analog Circuit from ±12 V Supply

=P1+P3+P4= 1153.25+1208.10+707.40 =3068.75 mW

Total Power for Analog Circuit

=P1+P2+P3+P4+P5= 5149.00+3068.75 =8217.75 mW

Total Power for Digital Circuit from +5 V Supply

=P6+P7+P8= 1485.00+3331.80+8070.00 =12886.80 mW

Total Power for the Card

=P1+P2+P3+P4+P5+P6+P7+P8= 8217.75+12886.80 =21104.55 mW
REPORT DOCUMENTATION PAGE Form Approved

OMB No. 0704-0188

Public reporting burden for this collection of information is estimated to average 1 hour per response, including the time for reviewing instructions, searching existing data

sources, gathering and maintaining the data needed, and completing and reviewing the collection of information. Send comments regarding this burden estimate or any other

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Washington, DC 20503.

1. AGENCY USE ONLY (Leave blank) 2. REPORT DATE

June 2000

3. REPORT TYPE AND DATES COVERED

Contractor Report

4. TITLE AND SUBTITLE

Advanced Atmospheric Water Vapor DIAL Detection System

5. FUNDING NUMBERS

PO L-68314D

6. AUTHOR(S)

Tamer F. Refaat and Hani E. Elsayed-Ali

WU 622-63-13-70

7. PERFORMING ORGANIZATION NAME(S) AND ADDRESS(ES)

 Old Dominion University

 Norfolk, VA 23508

 

 

8. PERFORMING ORGANIZATION

REPORT NUMBER

 

9. SPONSORING/MONITORING AGENCY NAME(S) AND ADDRESS(ES)

National Aeronautics and Space Administration

Langley Research Center

Hampton, VA 23681-2199

10. SPONSORING/MONITORING

AGENCY REPORT NUMBER

NASA/CR-2000-210301

11. SUPPLEMENTARY NOTES

Langley Technical Monitor: Russell J. DeYoung

12a. DISTRIBUTION/AVAILABILITY STATEMENT

Unclassified-Unlimited

Subject Category 33 Distribution: Nonstandard

Availability: NASA CASI (301) 621-0390

12b. DISTRIBUTION CODE

13. ABSTRACT (Maximum 200 words)

Measurement of atmospheric water vapor is very important for understanding the Earth's climate and water

cycle. The remote sensing Differential Absorption Lidar (DIAL) technique is a powerful method to perform such

measurement from aircraft and space. This thesis describes a new advanced detection system, which

incorporates major improvements regarding sensitivity and size. These improvements include a low noise

advanced avalanche photodiode detector, a custom analog circuit, a 14-bit digitizer, a microcontroller for on

board averaging and finally a fast computer interface. This thesis describes the design and validation of this new

water vapor DIAL detection system which was integrated onto a small Printed Circuit Board (PCB) with

minimal weight and power consumption. Comparing its measurements to an existing DIAL system for aerosol

and water vapor profiling validated the detection system.

14. SUBJECT TERMS

APD modelling; APD characterization; Digitizers; Lidar, DIAL

15. NUMBER OF PAGES

155

 16. PRICE CODE

A08

17. SEC U RITY CL ASSIF IC AT ION 

O F REPO R T

Unclassified

18. SEC U RITY CL ASSIF IC AT ION 

O F TH IS PA GE

Unclassified

19. SECURITY CLASSIFICATION

 OF ABSTRACT

Unclassified

20. LIMITATION

 OF ABSTRACT

 UL

NSN 7540-01-280-5500 Standard Form 298 (Rev. 2-89)

Prescribed by ANSI Std. Z-39-18

298-102