NBS Special Publication 705

Precision

Measurement

and

Calibration:

Electricity

A.O. McCoubrey - Editor

1985

U.S. DEPARTMENT OF COMMERCE

National Bureau of Standards
TM he National Bureau of Standards' was established by an act of Congress on March 3, 1901. The

Bureau's overall goal is to strengthen and advance the nation's science and technology and facilitate their effective application for public benefit. To this end, the Bureau conducts research and provides: (1) a

basis for the nation's physical measurement system, (2) scientific and technological services for industry and

government, (3) a technical basis for equity in trade, and (4) technical services to promote public safety.

The Bureau's technical work is performed by the National Measurement Laboratory, the National

Engineering Laboratory, the Institute for Computer Sciences and Technology, and the Institute for Materials

Science and Engineering

.

The National Measurement Laboratory

Provides the national system of physical and chemical measurement;

coordinates the system with measurement systems of other nations and

furnishes essential services leading to accurate and uniform physical and

chemical measurement throughout the Nation's scientific community, in- dustry, and commerce; provides advisory and research services to other Government agencies; conducts physical and chemical research; develops,

produces, and distributes Standard Reference Materials; and provides

calibration services. The Laboratory consists of the following centers:

• Basic Standards2 • Radiation Research

• Chemical Physics

• Analytical Chemistry

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Provides technology and technical services to the public and private sectors to address national needs and to solve national problems; conducts research in engineering and applied science in support of these efforts; builds and maintains competence in the necessary disciplines required to carry out this research and technical service; develops engineering data and measurement

capabilities; provides engineering measurement traceability services; develops

test methods and proposes engineering standards and code changes; develops

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the Federal Information Processing Standards Program, developing Federal ADP standards guidelines, and managing Federal participation in ADP

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Programming Science and

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Conducts research and provides measurements, data, standards, reference

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'Headquarters and Laboratories at Gaithersburg, MD, unless otherwise noted; mailing address Gaithersburg, MD 20899.

2 Some divisions within the center are located at Boulder, CO 80303. 'Located at Boulder, CO, with some elements at Gaithersburg, MD.
NATIONAL BUREAU OF STANDARDS

LIBRARY Precision Measurement and

Calibration: Electricity

Selected Papers on the Realization and

Maintenance of the Fundamental Electrical Units

and Related Topics

A.O. McCoubrey, Editor

Center for Basic Standards

National Measurement Laboratory

National Bureau of Standards

Gaithersburg, MD 20899

A compilation of previously published papers by the National Bureau of Standards including selected abstracts by non-NBS authors. This publication is a supplement to NBS Special Publication 300, Volume 3 (1968) and NBS

Handbook 77, Volume II (1961)

Issued October 1985

U.S. DEPARTMENT OF COMMERCE, Malcolm Baldnge, Secretary

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director
Library of Congress Catalog Card Number: 85-600580

National Bureau of Standards Special Publication 705

Natl. Bur. Stand. (U.S.), Spec. Publ. 705, 850 pages (Oct. 1985)

CODEN: XNBSAV

U.S. GOVERNMENT PRINTING OFFICE

WASHINGTON: 1985

For sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, DC 20402
FOREWORD

The requirements of industry and commerce for accurate measurements have

been the central focus of the National Bureau of Standards since the turn of

the century. The Bureau's research and development programs help incorporate

the advances of science and technology into the national measurement

standards of the United States. They also improve the methods for making

measurements, and permit NBS to refine its services — services that are essential to the use of measurement throughout the economy. Today, when the

country is spending an estimated $163 billion on measurement, the mission of

NBS takes on special significance. The Bureau's work is critical to the

nation as we respond to challenges of international trade in high technology

products and the demands of national defense.

That is why it is especially important that the results of NBS measurement-related programs be made available in a format that is both effective

and efficient for industry. Based upon past demand, the Precision Measurement and Calibration series of NBS publications has proven to be a valuable

tool for industrial metrologists . These are compilations of previously

published papers of NBS technical staff members, bringing together information that is widely scattered in the technical literature. Recent advances now require new compilations to serve those who depend upon NBS technology.

The latest volume is an important addition to this series.

Ernest Ambler

Director

iii

Editor»s Note

The Precision Measurement and Calibration Series of publications was initiated early in 1961 by the compilation, in three volumes, of approximately 150 previously published papers of the technical staff of the National

Bureau of Standards. The series was extended and brought up to date in the

years from 1968 through 1972 by the compilation, in ten volumes, of more than

500 additional previously published papers of the NBS staff. The demand for

these past editions of the Precision Measurement and Calibration Series and

the related sales provide convincing evidence for the usefulness of the com- pilations. They serve to collect the results of NBS programs that are otherwise scattered very widely throughout the technical literature. In the

collected form they are readily accessible to practical specialists in the

measurement standards laboratories of government and industry.

The present extension of the Precision Measurement and Calibration

Series in the field of Electricity has been undertaken in view of the past success of the series and continuing requests from industry for more publication of NBS work in a form that meets the needs of practicing metrologists

.

The new volume is mainly a compilation of papers published during the most

recent twelve years by the Electricity Division staff in the NBS Center for

Basic Standards. This collection of papers is, therefore, focused upon the

realization of the SI unit of electricity (the ampere) in the United States,

the practical maintenance of this unit in terms of the volt and the ohm and

topics concerned with the NBS calibration of related industrial electrical

measurement standards. While the compilation includes papers on the transfer

of d.c. units to a.c. measuring devices, it is by no means complete in terms

of NBS publications in the field of power frequencies, audio frequencies or radio frequencies. Many of the programs of the Electrosystems Division

(Gai thersburg) , the Electromagnetic Fields Division (Boulder) and the

Electromagnetics Technology Division (Boulder) are concerned with topics

that, as in the past, fall within a d.c. and low frequency classification;

however, it is no longer possible to include all the NBS publications from

these programs within the scope of a single volume. While it has been

necessary to limit the scope of this new volume primarily to papers published

by the staff of the NBS Electricity Division, an effort has been made to

address a wider range of interests by including, in appendices, the most

recent publications of bibliographies by the Electrosystems Division, the

Electromagnetic Fields Division and the Electromagnetic Technology Division.

An additional effort to extend the usefulness of this volume has been

made by including, as in the past, abstracts of closely related papers by

authors in other organizations. These abstracts reflect only a small fraction of such possibilities and no special NBS endorsement of the related publications is implied. Likewise, there is no implication that other equally

important publications do not exist.

v
The present compilation has been reviewed by a number of colleagues

within NBS and their comments have been incorporated. In this connection,

the Editor is particularly indebted to: Frank Hermach (Co-editor, with Ron

Dziuba, of Precision Measurement and Calibration; Electricity-Low Frequency,

issued December 1968), John Hastings, Robert Cutkosky, Barry Taylor and

Oscars Petersons. Their comments and suggestions all contributed substantially to the usefulness of the collection.

Draft copies of this compilation were also reviewed by a number of members of the National Conference of Standard Laboratories (NCSL) . All of the

comments and suggestions received from this group were very positive and

adoption of all the recommendations was only limited by the scope and size of

the compilation. It quickly became clear that papers on measurement assurance programs, as well as papers involving different regions of the frequency

spectrum and other areas of electrical technology must be considered as possible candidates for future volumes in the series.

Reviews by members of NCSL included recommendations for a number of

additions and deletions that contributed substantially to the practical value

of the work. In this connection it is a pleasure to acknowledge the contributions of Laurel Auxier and Arno Ehman of Beckman Instruments, Carl Quinn of

Simco Electronics, Fred Hume of Fluke, David Workman of Martin Marietta

Aerospace (Denver), Robert Weber and Klaus Jaeger of Lockheed Missiles and

Space, Graham Cameron of the Department of Defence (Canada), William Simmons

of Barrios Technology, Hartwell Keith and Gary Davidson of TRW, and Timothy

Driver of Sprague Electric.

Arthur McCoubrey, Editor
Table of Contents

Page

FOREWORD iii

EDITOR'S NOTE v

SURVEY PAPERS ON ELECTRICITY * 1

Impact of Quantized Hall Resistance on SI Electrical

Units and Fundamental Constants

B. N. Taylor (1985) 3

Report on the New NBS Determination of the Proton Gyromagnetic

Ratio

E. R. Williams, G. R. Jones, J. S. Song, W. D. Phillips, and

P. T. Olsen (1 985) 6

New Results from Previously Reported NBS Fundamental

Constant Determinations

B. N. Taylor (1984) 11

Electrical Units, Fundamental Constants, and the

1983 Least Squares Adjustment

B. N. Taylor (1984) 15

The National Measurement System for Electricity

Norman B. Belecki, Bernardine L. Dunfee, and

Oskars Peterson (1978) 23

Low Frequency Electrical Calibrations at the

National Bureau of Standards

F. L. Hermach (1967) 107

ABSOLUTE VALUES OF ELECTRICAL UNITS -I 115

Realization of Electrical Units: Classical Methods

The NBS Absolute Ampere Experiment

P. T. Olsen, V. E. Bower, W. D. Phillips, E. R. Williams,

and G. R. Jones, Jr. (1 984) 1 1 7

The Realization of the Ampere at NBS

Paul T. Olsen, Marvin E. Cage, William D. Phillips and

Edwin R. Williams (1980) 124

A Proposed Coil System for the Improved Realization of

the Absolute Ampere

P. T. Olsen, W. D. Phillips and E. R. Williams (1980) 128

New NBS Measurement of the Absolute Farad and Ohm

Robert D. Cutko3ky (1974) 144

vii
The Fundamental Electrical Standards: Present Status

and Prospects for Improvement

Robert D. Cutkosky (1971) 149

ABSOLUTE VALUES OF ELECTRICAL UNITS - II 157

Realization of Electrical Units: Quantum Methods

A Test of the Quantum Hall Effect as a Resistance

Standard

Marvin E. Cage, Ronald F. Dziuba and

Bruce F. Field (1984) 159

A High Accuracy Automated Resistance Bridge for

Measuring Quantum Hall Devices

Bruce F. Field (1984) 162

A Possible Quantum Hall Effect Resistance Standard

M. E. Cage, R. J. Dziuba and B. F. Field (1983) 165

The Quantum Hall Effect, I M. E. Cage and S. M. Girvin (1983) 170

The Quantum Hall Effect, II

S. M. Girvin and M. E. Cage (1983) 185

Progress Report on Investigations of the Quantum

Hall Effect as a Possible Resistance Standard

M. E. Cage, et al. (1983) 197

Laboratory Voltage Standard Based on 2e/h

Bruce Field and Victor W. Hesterman (1976) 208

Cryogenic Voltage Comparator System for

2e/h Measurements

Ronald F. Dziuba, Bruce F. Field and

Thomas E. Finnegan (1974) 211

Volt Maintenance at NBS via 2e/h: A New Definition

of the NBS Volt

B. F. Field, T. F. Finnegan and J. Toots (1973) 215

Summary of International Comparisons of

As-Maintained Units of Voltage and Units of 2e/h

Woodward G. Eicke, Jr. and Barry N. Taylor (1972) 227

A Flexible System with Two Selectable Ratios for

Use with Josephson Devices

David W. Braudaway (Abstract, 1978) 231

High Accuracy Potentiometers for Use with Ten

Millivolt Josephson Devices: II - Cascade

Interchange Comparator

A. Denenstein and T. F. Finnegan (Abstract, 1974) 231

viii
High Accuracy Potentiometers for Use with Ten

Millivolt Josephson Devices: I - Double SeriesParallel Exchange Comparator

T. F. Finnegan and A. Denenstein (Abstract, 1973) 232

AC Josephson Effect Determination of e/h:

A Standard of Electro-Chemical Potential Based on Macroscopic Quantum Phase Coherence in Superconductors

T. F. Finnegan, A. Denenstein and

D. N. Langenberg (Abstract, 1971 ) 233

VOLTAGE REFERENCE STANDARDS AND TRANSFER METHODS . . 235

A Sub-PPM Automated One-to-Ten Volt D.C. Measuring

SYSTEM

Bruce F. Field (1984) 237

An Automated Standard. Cell Comparator Controlled

by an Desk Calculator: A Preliminary Report

Robert E. Kleimann and Woodward G. Eicke, Jr. (1976) 241

A Resistive Standard for Measuring Direct

Voltages to 10 KV

Clifton B. Childers and Ronald F. Dziuba (1976) 247

A High-Resolution Prototype System for Automatic

Measurement of Standard Cell Voltage

David W. Braudaway and Robert E. Kleimann (1975) . 251

Standard Cell Enclosure with 20- yK Stability

Robert D. Cutkosky and Bruce F. Field (1974) 256

Standard Cell Calibration via Current Transfer

Edwin R. Williams, Paul T. Olsen and Bruce F. Field (1974) . . 260

The EMF-Temperature Coefficient of "Acid" Standard

Cells of the Saturated Cadmium Sulfate Type from

15 to 40 °C

Walter J. Hamer, Anna Skapars, and Bruce F. Field (1972) . . . 264

Thermodynamics of Standard Cells of the Saturated

Cadmium Sulfate Type

Walter J. Hamer (1971) 272

Designs for Surveillance of the Volt Maintained by

a Small Group of Saturated Standard Cells

W. G. Eicke and J. M. Cameron (1967) 293

Measurement Assurance Programs in a Field Environment

Woodward G. Eicke, Jr., Thomas F. Leedy, Brian R. Moore,

and Charles F. Brown (1982) 312

ix
Transfer of the Unit of Voltage

Norman B. Belecki (1968) 321

Regional Maintenance of the Volt Using NBS Volt

Transfer Techniques

Woodward G. Eicke, Jr. and Laurel M. Auxier (1974) 327

A Coordinated System of Maintaining and Disseminating

the Volt

Robert M. Shaw (Abstract, 1980) 332

The Response of Standard Cells to Alternating Currents

in the Frequency Range 10-1 0^ Hz

G. J. Slogett (Abstract, 1977) 332

Automatic Intercomparison of Standard Cells

Andrew F. Dunn (Abstract, 1 97 2*) 332

Internal Comparison of a Large Group of Standard Cells

J. Wilbur-Ham (Abstract, 1973) 333

Automatic Measuring System for a Control of Standard

Cells

Hiroyaki Hirayama and Yasushi Murayam (Abstract, 1972) .... 333

Maintenance of a Laboratory Unit of Voltage

Andrew F. Dunn (Abstract, 1971) 333

Truly Transportable Standard Cell Air Bath

David W. Braudaway (Abstract, 1970) 33^

The Application of the Direct Current Comparator

to a Seven Decade Potentiometer

Malcolm P. MacMartin and Norbert L. Kusters (Abstract, 1968) . 33^

The Construction and Characteristics of Standard Cells

George D. Vincent (Abstract, 1958) 335

RESISTANCE AND RESISTANCE APPARATUS 337

Automated NBS Measurement System

Kenneth R. Baker and Ronald F. Dziuba (1983) 339

A New Switching Technique for Binary Resistive

Dividers

Robert D. Cutkosky (1978) 3^

NBS Automated One-Ohm Resistance Measurements

Jerome J. Morrow (1976) 3^6

An Integrated System for the Precise Calibration of

Four-Terminal Standard Resistors

Thomas E. Wells and Earl F. Gard (1971 ) 350

x
Resistive Voltage-Ratio Standard and Measuring Circuit

Ronald J. Dziuba and Bernadine L. Dunfee (1970) 355

Evaluation of the NBS Unit of Resistance Based on

a Computable Capacitor

Robert D. Cutkosky (1961) 367

Alloys for Precision Resistors

C. Peterson (1954) 379

The Error Due to the Peltier Effect in DirectCurrent Measurements of Resistance

C. G. M. Kirby and M. J. Laubitz (Abstract, 1 973) 393

Resistance Comparisons at Nanovolt Levels Using

an Isolating Current Ratio Generator

L. Crovini and C. G. M. Kirby (Abstract, 1970) 393

CAPACITORS AND INDUCTORS 395

Transportable 1000 pF Standard for the NBS

Capacitance Measurement Assurance Program

George Free and Jerome Morrow (1982) 397

Testing to Quantify the Effects of Handling of

Gas Dielectric Standard Capacitors

Charles R. Levy (1982) 412

Losses in Electrode Surface Films in Gas Dielectric

Capacitors

Eddy So and John Q. Shields (1 979) 455

A Programmable Phase-Sensitive Detector for Automatic

Bridge Applications

Robert D. Cutkosky (1978) 461

Measurement of Four-Pair Admittances with Two-Pair

Bridges

John Q. Shields ( 1974) 463

Phase Angle Characteristics of Cross Capacitors

John Q. Shields (1 972) 471

Techniques for Comparing Four-Terminal-Pair

Admittance Standards

R. D. Cutkosky < 1 970) 475

A Varactor Null Detector for Audio Frequency

Capacitance Bridges

R. D. Cutkosky (1968) 491

Comparison Calibration of Inductive Voltage Dividers

Raymond V. Lisle and Thomas L. Zapf (1964) 499

xi
Designs for Temperature and Temperature Gradient

Compensated Capacitors Smaller Than Ten Picofarads

Robert D. Cutkosky (196*1) 504

A Technique for Extrapolating the 1 KC Values of

Secondary Capacitance Standards to Higher Frequencies

R. N. Jones (1963) ..... 507

Four-Terminal Pair Networks as Precision Admittance

and Impedance Standards

R. D. Cutkosky (1963) 521

Some Results on the Cross-Capacitances Per Unit Length

of Cylindrical Three-Terminal Capacitors with Thin

Dielectric Films on Their Electrodes

D. G. Lampard and R. D. Cutkosky (1960) 525

A Combined Transformer Bridge for Precise Comparison

of Inductance with Capacitance

Andrzej Muciek (Abstract, 1983) 533

AC/DC TRANSFER STANDARDS AND INSTRUMENTS 535

A Dual-Channel Comparator for AC-DC Difference

Measurements

Earl S. Williams and Joseph R. Kinard (1984) 537

An Automatic System for ac/dc Calibration

K. J. Lentner and Donald R. Flaeh (1983) 542

A..C. Voltage Calibrations for the 0.1 Hz to 10 Hz Frequency

Range

Howard K. Schoenwetter (1983) 548

The Practical Uses of ac-dc Transfer Instruments

Earl S. Williams (1982) 598

A Thermoelement Comparator for Automatic ac-dc

Difference Measurements

Earl S. Williams (1980) 631

An International Comparison of Thermal Converters

as ac-dc Transfer Standards

0. P. Galakhova, S. Harkness, Francis L. Hermaeh,

H. Hirayama, P. Martin, T. H. Rozdestvenskaya , and

Earl S. Williams (1980) 636

A Semiautomatic System for ac/dc Difference Calibration

Keith J. Lentner, Clifton B. Childers, and

Susan G. Tremaine (1980) 640

xii
An RMS Digital Voltmeter/Calibrator for Very-Low

Frequencies

Howard K. Schoenwetter (1978) 646

A Fast Response Low-Frequency Voltmeter

Bruce F. Field (1978) 656

AC-DC Comparators for Audio-Frequency Current and

Voltage Measurement of High Accuracy

Francis L. Hermach (1976) 661

Thermal Current Converters for Accurate ac Current

Measurements

Earl S. Williams (1 976) 667

An Investigation of Multij unction Thermal Converters

Francis L. Hermach and Donald R. Flach (1976) 672

A Low-Temperature Direct-Current Comparator Bridge

D. B. Sullivan and Ronald F. Dziuba (1974) 677

Thermal Voltage Converters and Comparator for Very

Accurate ac Voltage Measurements

E. S. Williams (1971 ) 682

Design Features of a Precision ac-dc Converter

Louis A. Marzetta and Donald R. Flach (1969) 692

APPENDIX I 701

Electrical and Electronic Metrology: A Bibliography

of NBS Electrosystems Division's Publications

(January 1968 through February 1984) 701

APPENDIX II 732

Metrology for Electromagnetic Technology:

A Bibliography of NBS Publications

(January 1970 through December 1983) 732

APPENDIX III 793

A Bibliography of the NBS Electromagnetic Fields

Division Publications

(January 1982 through December 1983) ..... 795

B Bibliography of the NBS Electromagnetic Fields

Division Publications

(January 1980 through December 1981) 808

C A Bibliography of Publications in the NBS Electromagnetic

Fields Division

(January 1970 through December 1979) 821

xiii

SURVEY PAPERS ON ELECTRICITY

Impact of Quantized Hall Resistance on SI Electrical

Units and Fundamental Constants

B. N. Taylor (1985) 3

Report on the New NBS Determination of the Proton Gyromagnetic

Ratio

E. R. Williams, G. R. Jones, J. S. Song, W. D. Phillips, and

P. T. Olsen (1985) 6

New Results from Previously Reported NBS Fundamental

Constant Determinations

B. N. Taylor (1984) 11

Electrical Units, Fundamental Constants, and the

1983 Least Squares Adjustment

B. N. Taylor (1984) 15

The National Measurement System for Electricity

Norman B. Belecki, Bernardine L. Dunfee, and

Oskars Peterson (1978) 23

Low Frequency Electrical Calibrations at the

National Bureau of Standards

F. L. Hermaeh (1967) 107

1

Metrologia 21, 37-39 (1985) metrologia

© Springer-Verlag 1985

Letter to the Editor

Impact of Quantized Hall Resistance on SI Electrical Units

and Fundamental Constants*

B. N. Taylor

Electricity Division, National Bureau of Standards, Gaithersburg, MD 20899, USA

Received: November 7, 1984

Abstract

With the discovery of the quantum Hall effect the SI

units ampere, volt, and ohm may be realized with

high accuracy by performing four fundamental-constant determinations requiring only a consistent set of

laboratory electrical units. Accurate values in SI units for most other constants, including the Avogadro and

fine-structure constants, may also be obtained from

the same measurements.

In two previous publications in this journal [1, 2] the author emphasized the important role that the

fundamental constants can play in realizing the

Systeme International d'Unites (SI) unit of current, the

ampere, .and how a direct measurement of the

ampere can provide an accurate, indirect value for the Avogadro constant. With the discovery of the

quantum Hall effect [3] the possibilities for realizing SI electrical units via fundamental-constant determinations has significantly increased and it is the

purpose of this note to briefly summarize these

possibilities. In particular, it will be shown how the

determination of four electric unit-dependent constants measured only in terms of a consistent set of

laboratory electrical units can be used to realize with

high accuracy the SI units ampere, volt, and ohm, and

at the same time provide accurate values in SI units

for most other fundamental constants including the

Avogadro constant NA and the fine-structure constant a. The four constants are the quantized Hall resis- tance RH = hie2 (h is the Planck constant and e the

elementary charge), the Josephson frequency-voltage

ratio lelh, the proton gyromagnetic ratio obtained by

the low-field method yp (low) (the prime means that the protons are in a spherical sample of pure H2 0 at 25 °C), and the Faraday constant F. (It is assumed

* Official contribution of the National Bureau of Standards, not

subject to copyright in the United States

throughout this note that the quantized Hall resis- tance is a true measure of hie2

, i.e., that any

corrections are negligibly small.)

A number of other constants are needed in addition to these four but they are either defined

numbers or electric-unit-independent constants which are well known. They are the permeability of vacuum ^ = An x 1CT 7 H/m, the speed of light in vacuum c = 299792458 m/s, the molar mass of the

proton Mp

, the Rydberg constant for infinite mass Rx , the magnetic moment of protons in H2 0 in units

of the Bohr magneton ju'p

/juB , and the ratio of the proton mass to the electron mass mJmt

. For simplicity we assume that Ru , lelh, y'(low),

and F are measured in the same laboratory and thus

in terms of the same laboratory electrical units. The

latter are taken to be VLAB defined in terms of the mean emf of a group of chemical cells, Qlab defined

in terms of the mean resistance of a group of resistors,

and ALAB = VLAB/QLAB . It is further assumed that all four quantities are measured at the same time so that

drifts in the laboratory units are of no consequence.

We may then define the following three quantities at a given time:

KA = ALAB /A Kv = VLAB /V KQ = CWft , where A, V, and Q are, of course, the SI units

ampere, volt, and ohm. In practice, determining the

ratios KA, Kw , and KQ is what is meant by realizing

the SI definitions of the ampere, volt, and ohm,

respectively.

From these definitions and the known relation- ships among the fundamental constants [4] one may

derive the following equations 1

:

1 Since yp (high)LAB = FLAB(/y>B) (wjp/me )/Mp

, where yp (high) is the proton gyromagnetic ratio determined by the high-field method, yp (high)LAB can be used in place of FLAB without

altering the principal conclusions of this note. However, a yp(high) experiment is more akin to the direct realization of the SI ampere via a force balance than is a Faraday-constant

experiment

3
KA =

Kv =

yp(iow)LAB

(m'p /Mb) (mp /me ) LAB

i : (1)

Table 1. Uncertainties of constants used to evaluate Eqs. (1)-(10)

. 16 -R-x (p'

p

/fii B ) (mp/me

)-

1/6

Kn = NA =

i _

yp(low)LAB (2 e/h)jAB (^h)lab Flab

' fipc

i

(fii'p

//4 B ) (2 e/h\AB

. 16 (#h)lab yP (low)LAB.

[(^h)lab (2 e/h)LAB FLAB]/2

' (M'M) (*h)lab (2 e/JiW 1 ' 3

_2/u0 R<» yP (low) LAB

1/3

(2)

(3)

(4)

(5)

where the subscript LAB indicates that these quantities are to be measured in terms of the appropriate

laboratory units. The latter are QLAB , ^lab» ^-lab

and ALAB for i? H , 2elh, yp (low) and F, respectively. It should be emphasized that none of Eqs. (l) — (5) contains an electric-unit-dependent constant which

must be measured in SI electrical units. In principle,

all that is required to define VLAB and Qlab and tnus

to determine (^h)lab> (2e/h) LAB , yp (low)LAB and

FLAB is a battery and a resistor!

It is also possible to derive equations involving

these same quantities from which one may obtain

indirect values in SI units for most other fundamental

constants of interest (except those involving the gas

constant). For example, we note the following

relations:

2 e/h = (2 e/h)LAB ATy

1

Rh = (^h)lab Ka = noca-*/2

e = 2 KA/[(RH) LAB (2 e/h)LAB}-

h = (/w0c/2)a

_ V

2 me = (Mp /NA )/(mp /me )

,

(6)

(7)

(8)

(9)

(10)

where all quantities are in SI units except those with

the subscript LAB and Kv , KQ , KA , a

-1 and NA are

to be taken from Eqs (1)— (5) as appropriate.

It is of interest to evaluate Eqs (1)— (10) using

existing measurements in order to see what levels of

uncertainty can be achieved with present-day measurement techniques. The National Bureau of Stan- dards (NBS) is the only laboratory to date which has

completed determinations of the four required elec- tric-unit-dependent constants [5—8]. However, the

experiments were not carried out at the same time but over the 9-year period 1975-1984. Although

Quantity Uncertainty3 (ppm)

Refs.

IK \ ^hJnbs A C\A1 Kl

V Z,C/ '*7NBS 0.031 l°J yp(iow)NBS 0.21 [7] ^NBS 1.33 [8, 10] ^0 defined

c defined Mp 0.012 [11, 10] R«, 0.0010 [12, 13, 10] 0.012 [14, 10] 0.043 [15]

All uncertainties are one-standard-deviation estimates. The

uncertainties of FNBS , Mp

, RK , and ju^Jju B are based in part on

analyses carried out by the author in connection with the 1983

least-squares adjustment of the fundamental constants [10]

has been maintained constant in time via the

Josephson effect with an estimated uncertainty of a few parts in 108 since 1972 July 1 [6], QNBS and thus

also ANBS may not be assumed to be time invariant

because Qnbs is defined in terms of the mean

resistance of five Thomas-type one-ohm resistors. Indeed, there is strong evidence that since the early 1970's QNBS has been decreasing at the rate of about

0.06 ppm/year [9]. Nevertheless, for illustrative purposes we assume that all four constants were

determined at the same time and in terms of the same

laboratory units. The one-standard-deviation uncertainties of the

four NBS measurements as well as those of the other

required constants are given in Table 1. These uncertainties, Eqs. (1)-(10), and a straightforward application of the law of error propagation then lead to the

results given in Table 2. The latter clearly show the

power of the fundamental-constant or indirect route

to the realization of SI electrical units; the uncertainties of these indirect values of KA , Kv , and KQ are comparable with or significantly less than the uncertainties which can be achieved by direct measurement. This statement applies as well to the indirect

values of the other constants listed in Table 2 in those

cases where direct measurements actually exist. In conclusion we point out that the four elec- tric-unit-dependent constants (^h)lab^ (2^)lab>

yp (low)LAB and FLAB need not be measured in the same laboratory if they can be reexpressed in terms of

the units of one particular laboratory as they existed

at a specific time. If, in fact, a number of laboratories use the Josephson and quantum Hall effects to define

their units of voltage and resistance and to maintain

them constant in time, then a measurement of

7p(low)LAB by any one of them at any time and of

4
Table 2. Uncertainties of quantities obtained from Eqs. (1) — (10) evaluated with constants having the uncertainties given in Table 1

^UallllL V

T Inpprta in tva

v

1 1 1 ^ ^ i tain L y

f nnm ^ VFF"v Ka 0.67 Kv 0.67 Kn 0.077 Na 1.33 a" 1 0.073

2e/h 0.67 Rh 0.073

0.68 h 1.34

1.33

a All uncertainties are one-standard-deviation estimates assuming

that the four constants (£h)nbs. (2<?//i)nbs> yp(low)NBS , and FNBS

of Table 1 were measured at the same time and in terms of the same laboratory units

FLAB by any one of them at any time will enable all of

them to determine their respective values of KA , Kv and Kq. With this in mind we emphasize the potential

importance of an improved determination of FLAB . A

measurement of the latter with an uncertainty of

0.5 ppm, which seems feasible, would yield a value of

2e/h with about half this uncertainty and a value of NA

of about the same uncertainty. It will be most difficult for direct measurements of either of these constants

to achieve such accuracies.

Acknowledgements. Helpful discussions with B. F. Field and E. R.

Williams are gratefully acknowledged. The data analyses carried

out in connection with the 1983 least-squares adjustment of the fundamental constants were done in collaboration with E. R. Cohen.

References

1. B. N. Taylor: Metrologia 9. 21-23 (1973)

2. B. N. Taylor: Metrologia 12, 81-83 (1976)

3. K. v. Klitzing, G. Dorda, M. Pepper: Phys. Rev. Lett. 45, 494-497 (1980)

4. See for example Refs. [1-3] and also: B. N. Taylor, W. H.

Parker, D. N. Langenberg: Rev. Mod. Phys. 41, 375-496

(1969); E. R. Cohen, B. N. Taylor: J. Phys. Chem. Ref. Data

2, 663-734 (1973)

5. M. E. Cage, R. F. Dziuba, B. F. Field: IEEE Trans. Instrum. Meas. IM-34, July 1985 (to be published)

6. B. F. Field, T. F. Finnegan, J. Toots: Metrologia 9, 155—166

(1973)

7. E. R. Williams, P. T. Olsen: Phys. Rev. Lett. 42, 1575-1579

(1979)

8. V. E. Bower, R. S. Davis. J. Res. Natl. Bur. Stand. 85, 175-191 (1980); L. J. Powell, T. J. Murphy, J. W. Gramlich:

J. Res. Natl. Bur. Stand. 87, 9-19 (1982); V. E. Bower, R. S. Davis, T. J. Murphy, P. J. Paulsen, J. W. Gramlich, L. J. Powell: J. Res. Natl. Bur. Stand. 87, 21-22 (1982)

9. B. N. Taylor: J. Res. Natl. Bur. Stand, (to be published)

10. E. R. Cohen, B. N. Taylor: J. Phys. Chem. Ref. Data (to be

published)

11. A. H. Wapstra, G. Audi: Nucl. Phys. A (to be published)

12. S. R. Amin, C. D. Caldwell, W. Lichten: Phys. Rev. Lett. 47, 1234-1238 (1981)

13. J. E. M. Goldsmith, E. W. Weber, T. W. Hansch: Phys. Rev.

Lett. 41, 1525-1528 (1978)

14. W. D. Phillips, W. E. Cooke, D. Kleppner: Metrologia 13, 179-195 (1977)

15. R. S. Van Dyck, Jr., F. L. Moore, P. B. Schwinberg: Bull. Am. Phys. Soc. 28, 791 (1983); and private communication,

1983

5
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985 Report on the New NBS Determination of the Proton

Gyromagnetic Ratio

EDWIN R. WILLIAMS, GEORGE R. JONES, member, ieee, JUN-SHOU SONG, WILLIAM D. PHILLIPS,

and PAUL T. OLSEN, member, ieee

Abstract—Vie describe a new measurement of the proton gyromagnetic

ratio in H2 0, y'p

, now in progress at NBS, including the construction of a single layer precision solenoid, the measurement of its dimensions by an inductive technique, and our latest dimensional measurement

results. We also discuss other improvements made since our last y'p

determination.

I. Introduction

THE measurement of the proton gyromagnetic ratio in H2 0 by the weak field method involves measuring two

quantities. First, one measures the dimensions of a precision

solenoid and calculates the magnetic field produced when a known current is passed through the solenoid windings. Sec- ond, one measures the precession frequency of protons in a sample of H2 0 placed in this calculable field using standard

nuclear magnetic resonance (NMR) techniques. The ratio of

the NMR angular frequency to the calculated flux density is 7p(low). The dimensional measurements have always been the more difficult part of the determination.

In this paper we describe our present experiment which employs a newly constructed precisiop solenoid. We also discuss the major changes we have made in our previous experiment

[1 ] and why we expect these changes to improve our accuracy from 0.2 ppm to better than 0.05 ppm.

The major motivation for a new 7p (low) experiment is to help test basic physical theories via the fundamental constants.

In particular, the fine structure constant, a, can be obtained

from the Josephson effect measurement of 2e/h and 7p (low)

via the following equation:

1 _

(2e/^)NBS

AR^ (fi NBs/ft) Mb 7p(low)NBS

0)

Here c is the speed of light in vacuum ;R m is the Rydberg constant for infinite mass; £2 NBS/f2 is the ratio of the National Bureau of Standards (NBS) as-maintained ohm to the absolute

or SI (International System) ohm; jUp

/jUs is the magnetic moment of the proton in units of the Bohr magneton (throughManuscript received August 20, 1984. This work was supported in part by the U.S. Department of Energy and the U.S. Air Force Office of

Scientific Research.

E. R. Williams, G. R. Jones, W. D. Phillips, and P. T. Olsen are with

the National Bureau of Standards, Washington, DC.

J. S. Song is with the National Bureau of Standards, Washington, DC,

as a Guest Scientist from Chongqing University, People's Republic of

China.

out, the prime indicates protons in a spherical sample of pure H2 0 at 25 °C); (2e/ft)NBS is the ratio of twice the elementary charge to the Planck constant measured in terms of the NBS as-maintained volt using the ac Josephson effect; and

7p(1ow)n Bs is the proton gyromagnetic ratio measured in terms of the NBS as-maintained ampere. The accurate determination of 7p (low) has become even more important with the discovery of the quantum Hall effect. The pertinent equation to obtain a from the quantized Hall re- sistance Rff is

-i _

2 n-NBS (J} s n VKH)

jU0 C ft NBS (2)

where fi 0is the permeability of free space. Equations (1) and

(2) can be combined to eliminate the quantity nNBS /£2, yielding

a"3 =

1 _ /V (2e/ft)NBs 2/Jo^oo Mb 7p(low)NBS

bs- (3)

Equation (3) is obviously not independent of (2) but repre- sents an alternative path to obtain a from the Hall resistance. However, the cube root dependence of (3) helps reduce the uncertainty in a. Of course, all three results could be combined together to obtain a least squares value for a. It is clear that by measuring y'p and f2 NBS/f2 more accurately, we can improve our understanding of the ac Josephson effect and the quantum Hall effect, while testing quantum electrodynamics

which requires a for the evaluation of its theoretical expressions (e.g., for the anomalous moment of the electron). Since

the completion of our last y'p experiment three other experimental results have been published. (These experiments are reviewed in [2]). The National Physical Laboratory in England and the Mendelyeev Institute of Metrology in the USSR

have reported values of y'p

that are in complete disagreement

with our 1979 result (10-15 standard deviation discrepancies).

In contrast, the Chinese value obtained at the National Insti- tute of Metrology agrees with our value. A new determination

of y'p should shed light on these discrepancies.

The dominant source of uncertainty in our previous experiment was the measurement of the diameter of the solenoid. We attempted to eliminate the need for a diameter measure- ment by using a special distribution of currents [1] , but this technique does not work well on short solenoids. We have,

therefore, constructed a long solenoid that will allow us to eliminate the need to measure the diameter.

U.S. Government work not protected by U.S. copyright

6
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

II. Construction of a Long

Single-Layer Solenoid

We fabricated the new 2.2-m long 0.3-m diam, fused silica solenoid by lapping techniques developed specifically for precision solenoid construction at NBS by Curtis, Moon, and

Sparks in 1932 [3]. In about 1958, Driscoll perfected the technique while lapping the solenoid used in our previous de- termination. Because of the critical role of the solenoid in de- termining 7p(low), we feel it is of interest to summarize

briefly its fabrication. We started with a rough cylindrical form of fused silica (ac- tually magnetically purified silica sand fused together) 2.2 m

long, with a 25-mm wall thickness, manufactured by U.S. Fused Quartz Company. 1 The form was first ground into a cylinder using a large lathe on which we mounted a grinding wheel. A special lathe was then constructed for the lapping

process by cutting an old lathe in the middle and extending it to handle the 2.2-m cylinder. This was necessary because we

could not use the large lathe for the time consuming lapping

process. We lapped the cylinder with a lap that made contact on two generators of the cylinder ±45° from the top. This lap was constrained so that it would not rotate with the cylinder

but was free to sit on the top of the cylinder. With grit applied, the lap was gently guided up and down the cylinder.

Frequently the cylinder had to be cleaned and measurements

taken of the diameter along its length. The time spent lapping

various areas was adjusted so as to obtain a diameter constant

to a few micrometers. A slight ellipticity (~3 jum) was de- tected and corrected by painting selected areas where less lapping was desired. This cylinder was then mounted in the large lathe, and a helix of 24 threads/inch was ground 0.3-mm deep. It was then

returned to the lapping lathe where a special lap was used to smooth the helical thread. To make this lap we machined an

inside thread into a 1-cm thick brass ring such that this ring

could, in principle, be threaded onto the 0.3-m diam helix ground in the cylinder. The brass ring, which was 10-cm long, was then cut along the axial direction into short pieces 3-cm

wide. A total of 46 of the 3 cm X 10 cm lap segments was

spaced along four generators of the cylinder. Two generators were located at ±60° from the vertical, and two were at ±30°.

The segments were then epoxied to a lap form made of fused

silica, while the lap segments were held with their threads

seated securely in the grooves of the cylinder while spaced

along the four generators.

The final lapping then proceeded as grit was applied to the

rotating cylinder, and this 46 segment lap was driven back and

forth along the top of the cylinder by the helical groove. The

lapping process, of course, averages out the variation in the

pitch of the helical groove. However, the diameter must be

measured repeatedly and the lapping time adjusted at various

positions along the cylinder to keep the diameter constant. A

short lap made of 14 segments was used for correcting small

variations in the diameter. The final helical groove took on a

1 Mention of a manufacturer does not signify an endorsement by the

U.S. National Bureau of Standards, nor does it imply that the item is the only or best item for any purpose.

smooth, nearly sinusoidal shape, and the lapping was stopped

when a 0.8-mm wire would fit tightly in this sinusoidal groove. We then wound the solenoid by drawing wire through dies

directly onto the cylindrical form. We started with 1-mm

diam oxygen-free pure copper that was plated with 1 /jm of

gold. The final wire diameter of 0.8 mm was obtained after drawing through two dies. The newly constructed solenoid was then taken to the non- magnetic facility where the measurement of its dimensions and

NMR frequency will be completed without any further movemerit of the solenoid. A shower of an insulating liquid, Fluori- nert FC-43 manufactured by the 3M Corporation, is used

to cool the solenoid and control its temperature to 0.02 °C. The temperature coefficient of expansion of the fused silica is about 0.5 ppm/°C.

III. Measuring the Solenoid

The solenoid has 2100 turns spaced about 1 mm/turn, and

it is necessary to measure their position and changes in diameter to about 0.02 fim. We have constructed a new measuring

apparatus for this purpose, but the principles upon which it is based are the same as in our earlier effort [4] . The following

is a brief summation of the method.

The basic idea is to use a magnetic inductive technique to locate the current in the windings. Coils A and A' in Fig. 1 form a linear differential transformer which locates the axial position of the current injected into selected (or activated) turns of the solenoid. Coils A and A' are connected so that

their output voltages cancel when centered on these activated

turns, and a servo system locks the coil assembly to the null point with a precision of 0.02 jum. A mirror (corner cube

M') located in the center of the coil assembly is part of a laser interferometer system, with the reference mirror (M) located

in the center of a second linear differential transformer. We choose to activate 10 turns at one time, and under computer control move the current injector to any selected 10

turns until information about all 2100 turns of the solenoid is obtained. Coils B, B', and C form a radius-to-voltage trans- ducer which measures the variations in the radius of the in- jected current loop. The voltage induced in coil C is inversely

proportional to the radius of the activated turns. To detect

small changes, we first "buck out" most of the voltage in coil C by using the two coils B and B'. At the same.time that we

are bucking out the voltage in coil C, we are also increasing the

sensitivity of this three-coil system to changes in the radius R

of the solenoid because the voltage in coils B and B' increases when the radius R increases. This three-coil radius-to-voltage transducer is then calibrated by having a few turns of wire on

both ends of the solenoid that are 50 ;um larger at one end and

60 urn smaller at the other end.

In order to insure the valid measurement of the axial and

radial variations, we use 10 other coils on the probe to measure the motion of the probe in the other degrees of freedom.

Fig. 2 is a cross section of the probe in the plane of the in- jected current. The coils labeled V are wired in opposition, as

are those labeled H; they are used to measure the vertical and

horizontal displacements using the gradient of the magnetic

field produced by the activated turns. The coils VA are wired

7
WILLIAMS era/.: NEW NBS DETERMINATION

Fig. 1. Solenoid dimensional measurement system used to determine

the axial position and radial variations of the wires. The five coils A, A', B, B', and Care attached to a silica tube, T, and can be pushed

or pulled along the axis of the solenoid. Coils A and A' locate the

axial position of the injected current and coils B, B', and C form a diameter-to-voltage transducer. Mirrors M and M' are part of a laser interferometer. The fused silica straight edge is used to guide the five- coil probe and forms part of the vacuum chamber.

CURRENT INJECTOR

AND STRAIGHTEDGE

Fig. 2. Cross section of the pick-up probe and solenoid in the plane of

the activated loop on the solenoid. A corner cube, CC, is located at the center. The coils marked V measure the vertical displacement, H

the horizontal displacement, HA the horizontal angle, and VA the

vertical angle. To measure the injector displacement, current is in- jected into the coil ID instead of the solenoid and the voltage in coil IDP is detected.

as one coil at the center and oriented orthogonally to the field of the activated turns. This pair measures the vertical angle

(pitch), while HA measures the horizontal angle (yaw). The

coil labeled ID is used in combination with the coil IDP to detect the current injector displacement with respect to the probe. Using these 10 coils, corrections are applied to the

axial position and radial variation raw readings, in order to ac- count for the fact that the straight edge that controls the posi- tion of the probe is not perfectly straight and that the current

injector does not traverse the top of the solenoid perfectly.

The radial variations voltage signal depends quadratically on

the V, H, VA, and HA signals, and it is imperative that the probe is centered where this dependence is an extremum in order to reduce the effect of straight-edge imperfections.

\

n

l\

/

i

\

1

1

J

n

s\

—r**'

1/

a bc TIME —

Fig. 3. Injection current and resulting signal. Upper trace: current in- jected into a portion of the solenoid. Second trace: voltage observed

across the pick-up coils when the probe is displaced +250 Mm from

the center of the activated loop. Third trace: probe is centered. Bot- tom trace: probe is displaced -250 fim. Letters indicate various times along the oscilloscope trace.

Another source of systematic error is very important and its elimination of interest. If the injected current is a sinewave,

then an "end effect" error is introduced. Because this current induces a voltage in the other windings of the solenoid and be- cause there is some capacitance between turns, some current flows in these windings, thereby producing an error signal in the pick-up coils [5]. This effect varies inversely with the square of the frequency and increases as the injection position

approaches the end of the solenoid. We eliminate it by employing the special waveform for the injected current shown in the upper trace of Fig. 3. (This waveform is now generated by

a digital waveform generator.) The output of the detector

coil is just the derivative of the injected current. The second

trace of Fig. 3 shows this output and results when the probe is +250 /im off null; the last trace is the output for -250 jum off

null. In the next to last trace we are locked to the null. Using

a lock-in amplifier with a flat frequency response, we can look

at the integral of the output voltage from time a to time b,

minus the integral from b to c. This integral is proportional to the mutual inductance or total flux change in the pick-up coil. The transient signals, however, integrate to zero. The effec- tiveness of this method is demonstrated by either showing that the null position is independent of frequency or that it is inde- pendent of the phase of the lock-in. We find an error of less than 0.04 pirn for a phase shift of up to 90°. The entire system is automated with a PDP-1 1 computer and

CAMAC interface. The laser interferometer is a two-frequency,

Zeeman-split laser system manufactured by Hewlett-Packard,

and its frequency is measured against an iodine stabilized laser [6] . The entire interferometer is in a vacuum, so there is no

correction due to the index of refraction of air. The laser and

probe displacement are aligned by using split photo diodes. With these diodes we can keep the alignment error to less than

0.01 ppm.

Fig. 4 shows measurements of the new solenoid in the criti- cal central region (±600 mm). We are now concentrating on

this region because the uncertainty in the calculated field due

8
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 198S

°- 0

1. 1 _ 'i i III! "i

1 _ „

1 00 mm

hi

1 I 1 1 1 1

2 Ampere

2oQ<

li'

1

!

I ^ i In ft

-60C -400 -200 0 200 400 600 WIRE NUMBER

Fig. 4. Radial variations (bottom) in and axial position variations (top) in micrometers versus wire number (~1 mm/wire number). The

radial variations are with respect to the center wires. For the axial variations a uniform pitch of 1.058588936 mm/turn has been subtracted from the laser interferometer reading at each wire number. The error bars are meant to represent one standard deviation uncertainty estimates.

to the measurement uncertainty of the outer windings is sig- nificantly less. Current is injected into 10 turns at a time, so

that the data plotted is the average position of the ten turns. The axial variations are a measure of turn position after a uni- form pitch of 1.058588936 mm/turn has been subtracted from the laser readings of that position. The radial variations

are a plot of the difference in radius of the ten turns at the in- dicated axial position relative to the center ten turns. These measurements show that the current in the solenoid flows

within a few micrometers of where current in a perfect solenoid

would flow.

IV. The Diameter Problem

With the high accuracy of the radius variation and pitch measurements, the diameter measurement should be the limiting

factor in the experiment. The basic problem in determining

the diameter is that even if one makes a perfect measurement

of the position of the surface of the copper wire, it is still diffi- cult to know where the current flows in the wire. The tech- nique developed to eliminate the need for a diameter measure- ment is to find a coil geometry which produces a magnetic

field that is nearly independent of the average diameter. With

a long solenoid we are able to find a suitable configuration in which extra current can be added to selected turns near the end of the solenoid. If done properly, this, in effect, gives us

the same properties of an infinitely long solenoid; namely,

that the magnetic field is independent of diameter and is uni- form at the center. With the aid of a computer, we found a configuration where the sum of the radius-variation weighting factor for each turn approaches zero, and the second-,

fourth-, and sixth-order gradients are compensated. The result (see Fig. 5) is that our solenoid can produce a field that has

the same diameter dependence as a 1-km long solenoid (i.e., the uncertainty due to the diameter measurement is essentially

Fig. 5. Compensation scheme for obtaining an "infinite" solenoid for NMR measurements. By using five current sources connected as indi- cated, a magnetic field is produced that is both uniform to 2 parts in 108 over a 6-cm sphere and essentially independent of the average diameter of the solenoid. Leakage between the five current sources must be kept very low.

eliminated) and a field that is uniform to 2 parts in 10 8 over a 6-cm diam spherical volume at the center of the solenoid. (The

water sample for the NMR measurements need only be 4 cm in diameter.)

V. Other Improvements

There is a remote chance that leakage between two adjacent

turns will cause an error in the calculated field. For our single helix solenoid a resistance of 30 k£2 across the most critical wires will result in a 0.01 ppm error. The problem is that the low resistance of the winding makes such leakage difficult to

detect. The technique we have developed to avoid this problem is very simple: We inject a dc current (5 A) into each turn, one turn at a time. If there is leakage to an adjacent turn, a leakage current will flow in it. However, the finite resistance of that turn produces a small voltage drop that can be detected with a pair of potential leads connected to the neighboring

wire. A leakage of 30 k£2 gives a voltage of 3 X 10~8 V;and

using a sensitive null detector and an automated system, we

can test every wire. The other major problem that we plan to solve is the correc- tion for the magnetic susceptibility of the fused silica cylinder.

We have taken three core samples and from their measured

small susceptibilities have calculated that the correction to the magnetic field is very small (<0.01 ppm), but this assumes

that the form is homogeneous. To verify that it is in fact homogeneous, we must measure the susceptibility correction

directly. To this end, we have constructed a large (0.6 m diam

by 2.2 m long) solenoid wound on a fused-silica cylinder. We

will measure changes in the field produced by this large sole- noid as we lower our precision solenoid in and out of this

field. We must detect changes of 0.01 ppm to measure the sus- ceptibility correction to the same accuracy.

VI. Conclusion

We are presently measuring our new solenoid with an accu- racy of 0.05 ppm. We have a precision of 0.02 ppm and ex- pect to improve the accuracy of the measurements to this level. We plan to carry out the NMR part of the experiment within

the next four to six months. This should give us a 0.1 ppm

preliminary value of y'p

. We then plan to spend another year

9
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

testing for systematic effects before obtaining final results. Ultimately we hope to obtain a value of 7p(low)NBS with an

uncertainty of 0.05 ppm or less, which should yield a value of a with an uncertainty of about 0.02 ppm.

Acknowledgment

The authors would like to thank E. Muth and D. Trammell

for their dedication in helping to construct the solenoid; H. Layer who calibrated the laser; G. Barrett who helped analyze

data; K. Maruyama, an NBS Guest Scientist from the Tokyo

Institute of Technology, Japan, who designed the injector

probe, and F. Fickett, of NBS/Boulder, for measuring the sus- ceptibility of the solenoid form material.

References

[1] E. R. Williams and P. T. Olsen, "New measurement of the proton

gyromagnetic ratio and a derived value of the fine-structure constant accurate to a part in 10 7 ," Phys. Rev. Lett., vol. 42, pp. 1575-1579, June 1979.

[2] E. R. Williams, P. T. Olsen, and W. D. Phillips, "The proton gyro- magnetic ratio in H2 0-a problem in dimensional metrology," in Precision Measurement and Fundamental Constants II, Natl. Bur. Stand. (U.S.), Spec. Publ. 617 (USGPO, Washington, DC), B. N.

Taylor and W. D. Phillips, Eds., 1984, pp. 497-504.

[3] H. L. Curtis, C. Moon, and C. M. Sparks, "A determination of the

absolute Ohm, using an improved self inductor," / Res. Natl. Bur. Stand., vol. 21, pp. 375-423, Oct. 1938, and private communication with R. L. Driscoll. [4] E. R. Williams and P. T. Olsen, "A noncontacting magnetic pickup

probe for measuring the pitch of a precision solenoid," IEEE

Trans. Instrum. Meas., vol. IM-21, pp. 376-379, Nov. 1972.

[5] H. P. Layer, "A portable iodine stabilized helium-neon laser," IEEE Trans. Instrum. Meas., vol. IM-29, pp. 358-361, Dec. 1980.

10
[ournal of Research of the National Bureau of Standards

Volume 90, Number 2, March-April 1985

New Results

From Previously Reported

NBS Fundamental Constant Determinations

B. N. Taylor

National Bureau of Standards, Gaithersburg, MD 20899

Accepted: December 13, 1984 A new treatment of previously reported results of three electric-unit-dependent fundamental constant experiments carried out at NBS over the last decade or so yields accurate, indirect values in SI units for a number

of important quantities. These include the fine-structure constant a, the Avogadro constant 7VA , the Josephson

frequency-voltage ratio 2e/h, and the quantized Hall resistance RH =h/e 2.

Key words: Avogadro constant; Faraday constant; fine-structure constant; Josephson frequency-voltage ratio; proton gyromagnetic ratio; quantized Hall resistance.

1. Introduction

Over the last decade or so the National Bureau of

Standards (NBS) has reported the results of three high

accuracy fundamental constant determinations carried out in terms of NBS as-maintained electrical units. The

three quantities are the Faraday constant FNm deter- mined in 1975 using a silver-perchloric acid coulometer

[l]

1

; the low field gyromagnetic ratio of the proton

(7p.l)nbs measured in 1978 using the so-called weak or low field method (the prime means the protons are in a

spherical sample of pure H2 0 at 25 °C) [2]; and the quantized Hall resistance (^?h)nbs=(^/^

2 )nbs (h is the Planck

constant and e the elementary charge) determined in 1983-1984 [3]

2 using the quantum Hall effect [6] in GaAs-Alx Gai^ x As heterostructures. Here and throughAbout the Author: B. N. Taylor, a physicist, is chief

of the Electricity Division in the NBS Center for Basic Standards.

'Numbers in brackets indicate literature references.

JThe result reported in Ref. [3] supplants entirely that given in Ref.

[4] since the temperature dependence of RH, as described in Ref. [5], was not properly taken into account in the earlier work of Ref. [4], We

assume throughout this paper that the quantized Hall resistance is a legitimate measure of h/e 2

, i.e., that any corrections are negligibly

small.

out this paper the subscript NBS indicates NBS electrical units; lack of a subscript indicates Le Systeme International d'Unites (SI) units. The NBS as-maintained electrical units in terms of

which FNBS, (7p,l)nbs, and 0R h)nbs have been measured

are (VNBS/0NBS) = ANBS and fl NBS . Here VNBS is the NBS

volt maintained constant in time since 1972 July 1 using

the ac Josephson effect with an uncertainty of a few

parts in 108

, the value of the Josephson frequencyvoltage ratio 2e/h adopted for this purpose being [7]

(2e//r)NBS =483593.420 GHz/VNBS . (1)

This implies that the ratio A'v^V nbs/V', where V is the SI volt, is given by

^v=VNBS/V (2a) = (2e/h)NBS/(2e/h) (2b)

and may be assumed to be time invariant within a few

parts in 108

. The quantity fi NBS is the NBS ohm defined in terms of

the mean resistance of five wire-wound, one-ohm re- sistors of the Thomas-type. Because flNBS is based on

artifact standards, one must assume that it is a timevarying unit. This implies that the ratio

11
(3)

also varies with time. The NBS ampere ANBS is not separately maintained

but is defined in terms of the NBS ohm and volt: AN bs = Vnbs/Ombs- It thus follows that ANBS also varies

with time as does the ratio

tfA =ANBS/A

=KV/K(1 .

(4a)

(4b)

The actual NBS electrical units in terms of which

^nbs. (Tp.l)nbs, and (^ h )nbs were measured are, re- spectively, ANBS, AnBS, and fl NBS . Because ANBS and

fi NBs are time varying units as just discussed and the three experiments were carried out over a 10-year period, the three results cannot be readily combined to obtain values for other quantities such as the fine- structure constant a and Avogadro constant A^A . This is unfortunate since, had they been measured at the same

time, it would have been a straightforward procedure to derive accurate, indirect values in SI units for these constants as well as for most others of interest. The

relevant equations for doing so may be obtained from

the known relationships among the constants [8-11]; some of the more important expressions are:

a = (M-p/>b) R H(t)sBs(2e/h

2^-0^00 Tp.lXOnbs

NA = [R H(t hBS(2e /h ) NBSF(t ) NBS]/2

2e/h =

(5)

(6)

l62 Ri(^/nB )(m D/me )

}

Ho

4 c

6Mp

3

RH(0NBs(2e/h^BSF(t)^ BS 7p,l(' )nbs

RH = fXoCa

] /2 =R H(t)NBSKn(t)NBS

(7)

(8a)

where the time dependencies of (/? h)nbs. (7p,l)nbs, and

FNBS have been explicitly indicated [(2<?//z) NBS is time

independent since VNBS is defined in terms of it through

eq (1)]. In these expressions u.0 =47rX 10~ 7 H/m is the

permeability of vacuum; c =299792458 m/s is the speed

of light in vacuum; ju, p/u, B

is the magnetic moment of the proton in units of the Bohr magneton (0.012 parts-permillion or ppm current uncertainty 3 ); R x is the Rydberg

constant for infinite mass (0.0010 ppm current uncertainty); Mp is the molar mass of the proton (0.012 ppm

current uncertainty); and mv/me is the ratio of the pro- ton mass to the electron mass (0.043 ppm current uncertainty). In the past, these and related expressions have been

evaluated by assuming that fl NBS has been constant in time and by allowing an additional uncertainty of about

0.01 ppm per year for its possible drift [1,2,4]. However,

recent comparisons of fl NBS with the resistance units of

other national laboratories show that since about 1970

the NBS ohm has likely been decreasing by approximately 0.06 ppm/yr. In the course of reviewing the implications of such a drift on eqs (5-10), it became

apparent that the problem of the time variation of ft NBS could be neatly solved by making use of one other NBS

result, namely, the determination in 1973-1974 via the NBS calculable cross capacitor of the ratio A'n^fiNBs/ft

[12]. With the addition of this single measurement, the

drift rate of the NBS ohm may be uniquely determined,

the values of (R H ) NBS, (7p,l)nbs, and FNBS converted to the same measurement date, and eqs (5-10) and their numerous extensions readily evaluated. The only crit- ical assumption required is that the drift of the NBS ohm

has been linear since the time of the calculable capacitor

determination of Kn . However, this is supported by the observed linear time dependencies of the measured dif- ferences between each of the five resistors which comprise 0NBS and their mean [13]. We now briefly summarize how the calculation proceeds. The linear drift rate assumption enables one to write

Ka(t ) sfl(/ ) NBS/n

km[l +b(t - /„)], (ii)

F =

Hoc\ix'p/>b) RH(t)NBS(2e/h ) NBS \6R

(/A

p/u. B )(m p/mc ) Tp.iXOnbs/^Onbs

Kn(0 =

\j&\ix'v/>B) (2e//z)NBS

16/? x/? h (0nbs7;.l(0nbs.

(8b)

(9)

(10)

where t is the time in years measured from some arbi- trary date, ta is the mean date of the NBS calculable

capacitor experiment with Kn {tn) the mean value obtained, and b is the relative drift rate of fi NBS . 4 It then

Throughout, all uncertainties are one standard deviation estimates.

4The time period over which Kn was measured was sufficiently short and the random scatter sufficiently large that the effect of the

drift of Unbs was indiscernible and hence negligible. This is also true of the measurements of FNBs. (7p.l)nbs, and (/? h )nbs12
listed in tables 1 and 2, to which these comments apply:

Table 1 . The values quoted for the nonexact constants are based on the results reported in the original

references and (with the exception of mv/me) analyses

carried out by the author in connection with the 1985

least squares adjustment of the fundamental constants

[14]. However, these analyses have generally led to only

minor changes in the original results. Table 2. Only the value of FNBS has been at all changed from that originally reported. The 0.2 ppm net increase arises from a number of positive and negative

corrections. New measurements of the Au and Ta impurity content of the silver used in the experiment [20] were specifically undertaken as a result of the author's

reanalysis of the experiment for the 1985 adjustment. It should be noted that the 0.031 ppm uncertainty assigned

(2e/h)NBS is an estimate of how well the NBS Josephson

effect voltage standard apparatus implements the defini- tion of VNBS [see eq (1)]. The uncertainties associated

with relating the working voltage standards used in the

(Tp.l)nbs and FNBS experiments to the voltage standards

used to preserve or store VNBS between Josephson effect calibrations are included in the uncertainties assigned

FNBS and (y^W as given in the table. The results of the calculations are:

cr' = 137.035981(12) (0.089 ppm) (15)

/?h = 25812.8041(23) Cl (0.089 ppm) (16)

2e/h -483597.91(32) GHz-V" 1 (0.67 ppm) (17)

NA = 6.022 1438(80) X 1023 mol" 1 (1.33 ppm) (18)

F = 96485.381(65) A-s-mol" 1 (0.67 ppm), (19)

Table 2. Summary of NBS electric-unit-dependent data used to evaluate eqs (5-10).

Quantity Mean date of measurementValue' Uncertainty

(ppm)

Refs.

(2e/A)NBS Continuous since 1972 July 1483593.420(15) 0.031 [7]

•Fnbs 1975 March 15 96486.19(13) 1.33 [1.20,14]

(7p,l)nbs 1978 March 22 2.67513229(57) 0.21 [2]

(•^h)nbs 1983 November 27 25812.8420(12) 0.047 [3]

1973 December 2 1 -0.819(27) x lO" 6 0.027 [12]

'The units for (2e/A)NBS experiment the NBS tesla Tare GHz VN

"

B

's ; for FNBS, NBS a ANBS. ANBS s-mol-'; for (yp, L)NBS. '°8 s ~

'•Tnb's; for (R H ) NBS , nNBS . Note that in the(7p L ) NBS

follows from this equation and the way fi NBS enters the determination of FNBS, (7p,l)nbs, and CR h)nbs that

F(t)vBs=F(tF) NBS [l+b(t-tF)] (12)

7PxC )NBS = 7;. L('y)NBs[ 1—A(f — ty)] (13)

*h(0nbs = *h('r)nbs[1 -b(t-tR )], (14)

where tF, ty

, and t R are the mean dates of the Faraday

constant, proton gyromagnetic ratio, and quantized Hall

resistance determinations, respectively, and .F(?f)nbs. 7p,l(^)nbs. and /? h ('r)nbs are the mean values obtained.

The drift rate b may then be calculated by substituting

eq (13) for yp,L(0NBs into eq (10) and equating the result to eq (1 1) with t = /R. Once b is in hand, eqs (5-10) may

be evaluated with the aid of eqs (12-14). Of course, the law of error propagation must be strictly obeyed in or- der to obtain correct uncertainties for the calculated

quantities.

We have carried out such calculations with the data

Table 1. Summary of defined and electric-unit-independent constants used to evaluate eqs (5-10).

Quantity Numerical Uncertainty Refs. value 1 (ppm)

4ttX10- 7 defined

c 299792458 defined 0.001520993127(18) 0.012 [15,14] 10973731.529(11) 0.00!0 [16,17,14] A/p 0.001007276470(12) 0.012 [18,14] 1836.152470(79) 0.043 [19]

'The units for fio are H-m -1; for c, m-s~ '; for R x , m" 1

; for A/p, kg mol -1

.

13
with b =(-0.0650±0.0102) ppm/yr. 5 The value for Kn on 1985 January 1 is 1 — ( 1 .539 ±0. 107) ppm. While it is not the purpose of this paper to make detailed comparisons of these results with others obtained by either

direct or indirect means, we do note that in most cases where other values exist the agreement is statistically

acceptable. We further point out that the value of 2e/h

derived here is (9.29 ±0.67) ppm larger than the value

used to define VNBS, implying that VNBS is (9.29±0.67)

ppm smaller than the SI volt [see eq (2b)]. This value of

2e/h is also (8.09±0.67) ppm larger than the value

483594 GHz/V which is in use in many other national

laboratories and which was recommended by the Comite Consultatif d'Electricite in 1972 [23]. In closing we emphasize that the values given here for the various constants, in particular the fine-structure

constant, are the best that can be obtained based solely on

existing NBS electrical measurements . 6 If the three quantities (7p, l)nbs» (-Rh)nbs. and Kn had been determined at the same time, then any two would have been sufficient to determine a~\ For example, eq (5) yields a value of

a" 1 from (y^OwBs and (./?h)nbs- The comparable equations for the two other pairs of measurements are

7

(20) and the quantum Hall effect value derived from eq

(21) , perhaps the value given here should simply be

referred to as the NBS condensed matter value.

a =

c(u.;/u.B) (2e/h)NBS AR , A:n(7;, L) NBs

a

l = 2(i? H)NBsA'n//Xo<:.

(20)

(21)

By treating the drift rate of the NBS ohm as a variable we remove the redundancy or overdetermination inherent in the three measurements and obtain a unique value

for a" 1 and all other quantities. Since this approach

eliminates the distinction between the so-called Jo- sephson effect value of a

-1 traditionally derived from eq

5 This value is in agreement with b = —(0.0562 ±0.0048) ppm/yr

which the author derived from a linear least squares fit to the results of direct and indirect comparisons of Unbs with the unit of resistance Hnml maintained constant in time by Australia's National Measurement Laboratory (NML) with the NML calculable cross capacitor [21]. We also note that the Bureau International des Poids et Mesures unit of resistance IIbipm is based on the mean of six resistors, two of which are of the same type used to define Unbs- Comparisons

of Hbipm with Onml dating back to 1964 show that Ubipm varies quite

linearly with time. Since the time dependencies of the measured differ- ences between Hbipm and each of the two NBS resistors which partially define fi B iPM are also observed to be linear [22], it may be con- cluded that the linear drift rate assumption for fl NBs is justified. If in fact one assumes the existence of a quadratic component, any reasonable estimate of its magnitude is sufficiently small that its effect is inconsequential.

6 "Best" or "recommended" values in the traditional sense would, of

course, require taking into account the relevant data available from all sources. 'Note that eliminating Kn from eqs (20) and (21) yields eq (5) while

equating the two yields eq (10).

I gratefully acknowledge helpful discussions with B. F. Field and E. R. Williams and the assistance of V. E. Bower, R. S. Davis, E. R. Williams, P. T. Olsen, M. E. Cage, R. F. Dziuba, and B. F. Field in reviewing their experiments. The data analyses carried out in connection with the 1985 least squares adjustment of the fundamental constants were performed in collaboration

with E. R. Cohen.

References

[1] Bower, V. E., and R. S. Davis, J. Res. Natl. Bur. Stand. 85, 175 (1980)/Powell, L. J.; T. J. Murphy and J. W. Gramlich, J. Res. Natl. Bur. Stand. 87, 9 (1982)/Bower, V. E.; R. S. Davis, T. J. Murphy, P. J. Paulsen, J. W. Gramlich, and L.

J. Powell, J. Res. Natl. Bur. Stand. 87, 21 (1982).

[2] Williams, E. R., and P. T. Olsen, Phys. Rev. Lett. 42, 1575

(1979).

[3] Cage, M. E.; R. F. Dziuba and B. F. Field, IEEE Trans. In- strum. Meas. IM-34, June 1985, to be published.

[4] Tsui, D. C; A. C. Gossard, B. F. Field, M. E. Cage, and R. F. Dziuba, Phys. Rev. Lett. 48, 3 (1982).

[5] Cage, M. E.; B. F. Field, R. F. Dziuba, S. M. Girvin, A. C. Gossard, and D. C. Tsui, Phys. Rev. B 30, 2286 (1984).

[6] Klitzing, K.V.; G. Dorda and M. Pepper, Phys. Rev. Lett. 45, 494 (1980).

[7] Field, B. F.; T. F. Finnegan and J. Toots, Metrologia 9, 155

(1973).

[8] Taylor, B. N., IEEE Trans. Instrum. Meas. IM-34, June 1985, to be published.

[9] Taylor, B. N., Metrologia 12, 81 (1976); Metrologia 9, 21 (1973).

[10] Cohen, E. R., and B. N. Taylor, J. Phys. Chem. Ref. Data 2, 663

(1973).

[1 1] Taylor, B. N.; D. N. Langenberg and W. H. Parker, Rev. Mod.

Phys. 41, 375 (1969).

[12] Cutkosky, R. D., IEEE Trans. Instrum. Meas. IM-23, 305

(1974). [13] Dziuba, R. F., private communication (1984).

[14] Cohen, E. R., and B. N. Taylor, J. Phys. Chem. Ref. Data, to be

published.

[15] Phillips, W. D.; W. E. Cooke and D. Kleppner, Metrologia 13, 179 (1977).

[16] Amin, S. R.; C. D. Caldwell and W. Lichten, Phys. Rev. Lett.

47, 1234 (1981).

[17] Goldsmith, J. E. M.; E. W. Weber and T. W. Hansen, Phys. Rev. Lett. 41, 1525 (1978).

[18] Wapstra, A. H., and G. Audi, Nucl. Phys. A, to be published.

[19] Van Dyck, R. S. Jr.; F. L. Moore and P. B. Schwinberg, Bull. Am. Phys. Soc. 28, 791 (1983) and private communication

(1983).

[20] Greenberg, R. R., private communication (1984).

[21] Thompson, A. M., Metrologia 4, 1 (1968)/Clothier, W. K., Metrologia 1, 35 (1965).

[22] Leclerc, G., and T. J. Witt, private communication (1984).

[23] Com. Int. Poids Mes., Com. Consult. d'Electricite Trav. 13c Session (1972) p. E-13.

14
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

Electrical Units, Fundamental Constants, and the 1983

Least Squares Adjustment

BARRY N. TAYLOR, fellow, ieee

Abstract—Ihis review touches upon four topics: (1) The International System or SI electrical units, specifically, the volt (V), ohm (fi), ampere

(A), and the so-called laboratory or as-maintained units for the same

quantities; (2) the relationships between these laboratory units, experi- ments to realize their SI definitions, and the fundamental constants of

nature: (3) the 1983 least squares adjustment of the constants; and (4) future electrical measurements research which can improve our knowledge of the constants.

I. Units

THIS section reviews the International System or SI defini- tions of the SI base electrical unit, the ampere, and the two most often used derived electrical units, the volt and ohm;

how in practice these three units are represented in the laboratory; and how these practical laboratory representations are maintained.

A. SI Electrical Units

The ampere is related to the three SI base mechanical units, the second, meter, and kilogram [1] , [2] . The second is de- fined as the duration of 9 192 631 770 periods of the radiation corresponding to the transition between the two hyperfine

levels of the ground state of the cesium 133 atom. The meter is- defined as the length of the path travelled by

light in vacuum in 1 /299 792 458 of a second. This is equivalent

to adopting an exact value for the speed of light in vacuum c, namely,

c= 299 792 458 m/s. (1)

The kilogram, the only SI unit still based on an artifact, is defined as the mass of the international prototype of the kilo- gram kept at the International Bureau of Weights and Measures (BIPM) in Sevres, France (a suburb of Paris). The SI unit of force, the newton (N), is derived from these

three base mechanical units using Newton's second law; it is defined as that force which gives to a mass of one kilogram an

acceleration of one meter per second, per second. Thus 1 N =

1 kg m • s~

2

. The SI unit of energy, the joule (symbol J), is derived from the newton and meter and is defined as the work

done when the point of application of a 1-N force moves a dis- tance of 1 m in the direction of the force. Hence 1 J = 1 N • m.

Finally, the SI unit of power, the watt (W), is derived from the

joule and the second, and is defined as the power which in one

second gives rise to energy of one joule. Thus 1 W = 1 J • s -1

.

Manuscript received September 26, 1984. The author is with the Electricity Division, National Bureau of Standards. Gaithersburg, MD 20899.

The ampere is then defined as that constant current which, if maintained in two straight parallel conductors of infinite length,

of negligible cross section, and placed one meter apart in vac- uum, would produce between these conductors a force equal

to 2 X 1CT 7 N/m of length. This definition implies via the

Biot-Savart law that the magnetic permeability of vacuum n0

is an exact constant given by

Mo =4ttX 10" 7 N/A 2

. (2)

The SI units for potential difference (electromotive force)

and electric resistance, the volt (V) and the ohm (f2), respectively, are then derived from the watt (and thus the three base

mechanical units) and from the base electrical unit, the ampere,

as follows: The volt is the difference of electric potential between two

points of a conducting wire carrying a constant current of

1 A, when the power dissipated between these points is equal

to 1 W.

The ohm is the electric resistance between two points of a conductor when a constant potential difference of 1 V, applied to these points, produces in the conductor a current of

1 A, the conductor not being the seat of any electromotive

force.

B. Laboratory or As-Maintained Electrical Units Because the SI definitions of the volt (V), ohm (ST), and

ampere (A) are difficult to realize in practice, national standards

laboratories (such as the National Bureau of Standards in the

U.S.) have historically used practical representations of them

based on artifacts. For example, a group of electrochemical standard cells of the Weston type (each with an EMF of order

1.018 V) may serve to define the laboratory or as-maintained

unit of voltage J^ab* and a group of precision wire-wound

resistors of the Thomas or similar type (each with a resistance of order 1 £2) may serve to define the laboratory or as-maintained unit of resistance nLAB V>\ • The laboratory unit of

current A LAB is then defined in terms of I^ab and ^lab

via ^lab = *1ab/^lab (3)

and does not have its own separate representation.

Such an as-maintained system of electrical units immediately

raises two questions: 1) How does one ensure that VLAB and

^lab are contant in time, based as they are on artifacts? 2)

How does one ensure that JIab and ^lab are consistent with

U.S. Government work not protected by U.S. copyright

15
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE I98S

their SI definitions? Problem 1 introduces the idea of monitoring or maintaining laboratory units while Problem 2 introduces

the idea of carrying out absolute realizations of the SI electrical units. Ideally, one would like to solve both problems simulta- neously, that is, maintain a unit constant in time and consis- tent with its SI definition by the same means, at the same time,

and at a level of accuracy which is in keeping with the inherent

stability of the artifacts. However, because this stability is at the level of parts in 10 7 per year (or for some very well-aged

cells and resistors at the level of parts in 10 8 per year), it has not yet proved feasible to do so in most laboratories. The two

problems are, therefore, considered separately.

C. Maintaining Laboratory Units

Most national laboratories now use the ac Josephson effect

in superconductors to maintain their unit of voltage. For

our purposes a Josephson device may be viewed as a perfect

frequency-to-voltage converter, the constant of proportionality

being the invariant fundamental constant ratio 2e\h (e is the ele- mentary charge, h is the Planck constant). Because frequencies can be readily measured to very high accuracy (parts in 10' 3 if necessary), the Josephson effect can be used to define and maintain VhAB to an accuracy limited only by the accuracy with

which the voltage across the Josephson device can be compared

with the EMF of a standard cell. Typically this is in the range 0.01-0.1 parts-per-million (symbol ppm). (Throughout this paper all uncertainties are meant to correspond to one standard

deviation estimates.) The defining equation for VLA% is

(2 e/h)LAB = 483 59? GHz/ VLAB (4)

where {2ejh\AB is the specific value of 2ejh adopted by the laboratory to define Plab- (The question mark is meant to

indicate that several different values are in use at the various

national laboratories; 483 594 is the most common numerical

value.) The standard cell now serves only as a "flywheel," that

is, as a means of preserving or storing KLAB between Josephson

effect measurements.

While not yet in as wide a use as the Josephson effect, the quantum Hall effect (QHE) [4] promises to do for resistance- unit maintenance what the Josephson effect has done for volt- age-unit maintenance. For our purposes a QHE device can be

viewed as a resistor the resistance of which depends only on

the fundamental constant ratio h/e2. Thus the QHE, which

manifests itself in certain semiconductor devices when cooled

to cryogenic temperatures and in the presence of a large magnetic field, can be used to define and maintain £2LAB t0 an

accuracy limited only by the accuracy with which the resis- tance of the device can be compared with the resistance of a

l-f2 resistor. Eventually this is expected to be in the range

0.01 to 0.1 ppm for all laboratories; 0.05 ppm is the smallest uncertainty reported to date. The defining equation for f2]_AB

is

(h/e 2

)LAB = 25 812.8? f2 LAB (5)

where, by analogy with (4), (h/e 2 ) LAB is the specific value of

h/e2 adopted by the laboratory to define £2 LAB . (In this case the question mark means that a specific value has yet to be

adopted by any laboratory.) The combination of constants

h/e2 has been termed the quantum Hall resistance Rfj. It is related to the inverse fine-structure constant a

-1 (which is =137), the dimensionless coupling constant or fundamental

parameter of quantum electrodynamic theory (QED), by

h/e2 = RH = n0 ca

l l2. (6)

Since ju 0 and c are exact constants, if a

-1

is known with a given

uncertainty, Rfj will be known with the same uncertainty. By

analogy with the standard cell, the l-f2 resistor now serves only

to store ^2lab between QHE measurements.

II. Absolute Realizations and

Fundamental Constants

While the Josephson and quantum Hall effects allow VLAB

and Olab to be maintained constant in time to between

0.01 and 0.1 ppm, thereby solving Problem 1 of Section I-B, unless 2e\h and RH are known in SI units to the same level of

accuracy, the as-maintained and SI systems of electrical units

will not be consistent and Problem 2 will remain unsolved. In practice there are two general approaches to determining 2e/h

and RH in SI units. The first is to carry out experiments to

realize directly the SI definitions of V, £2, and A; the second is to carry out experiments to determine various fundamental

constants from which V, £7, and A may be indirectly derived. As will shortly be shown, at the present time the latter ap- proach often yields smaller uncertainties than the former.

A. Absolute Realizations: Direct

Currently, direct absolute realizations of the SI electrical units are based on experiments which use electromagnetic

theory in combination with the SI definitions. Experiments to

realize the SI electrical units as they are actually defined (see

Section I-A) are as yet too imprecise to be of interest. The fol- lowing is a brief summary of how the SI volt, ohm, and

ampere are being realized directly at present.

1) Volt: By measuring the force between electrodes to which a voltage known in terms of l^AB has been applied, and

the capacitance between the electrodes (determined in SI units

via a calculable cross-capacitor), the ratio VLAB/V can be obtained. In practice, determining this ratio is what is meant by

the terms "absolute realization of the volt" or "realization of

the SI volt." Defining (ATv )lab = ^lab/v and recognizing

that (2e/h)LAB is used to define VLAB , we may write

(Kv)hAB =VlAB/V = (2elh)LAB l(2e/h). (7)

A realization of the SI volt is thus equivalent to determining

2e/h in SI units (recall that (2e/h)LAB is an adopted number

with no uncertainty). The smallest uncertainty reported to

date for such an experiment is 2.4 ppm, some two orders of

magnitude larger than the uncertainty with which VLAB may

be maintained via the Josephson effect. 2) Ohm : The SI ohm may be realized using a so-called Thompson-Lampard calculable cross- capacitor which allows

the capacitance of a special configuration of electrodes to be

calculated (in SI units) to very high accuracy from a single

length measurement and the permittivity of vacuum, e 0

- (Since e0 = 1/MoC

2 and ju 0 and c are exactly known constants,

e0

is also exactly known.) The impedance of the calculable

16
TAYLOR: ELECTRICAL UNITS, FUNDAMENTAL CONSTANTS

capacitor is compared with the resistance standards which

represent £2lab using a complex series of bridges. Defining

(^o)lab - ^lab/^ and recognizing that (Rh)lab is used to

define ft LAB , we may write

CK"n)LAB = £2LAB /ft - RhI(Rh)lab- (8)

A realization of the SI ohm is thus equivalent to determining

RH in SI units, which in turn is equivalent to determining the

inverse fine-structure constant [see (6)] . The uncertainty in such an experiment is on the order of 0.05-0.1 ppm, comparable to the yearly stability of good resistance standards and the accuracy with which f2 LAB may be currently maintained via RH. However, because of the complexity of the ex- periment only one laboratory in the world has yet been able to maintain their laboratory unit of resistance consistently via the calculable capacitor. It would thus appear that the QHE

will become the method of choice in most laboratories. 3) Ampere: By measuring the force between coils of known

dimensions carrying a current known in terms of ^4lab> tne

ratio A LAB/A can be determined. Defining (Xa)lab = A lab/A, we may write

(Xa)lab lab/A- - C^v )lab /C^n )lab • (9)

A realization of the SI ampere measures neither 2e/h or RH

but the product (2elh)RH [see (7) and (8)] , which is equivalent

to measuring the elementary charge e in SI units. The smallest uncertainty reported to date for such an experiment is 4 ppm,

about two orders of magnitude larger than the uncertainty

with which A^AB can be maintained via the Josephson and

quantum Hall effects.

B. Absolute Realizations: Indirect As indicated above, (Ky ) LAB ,(Kn)hAB , and (A:A )LAB may

also be obtained from appropriate combinations of various fundamental physical constants. The constants required, in addition to the exactly defined quantities [jl 0 , c, (2e/h)hAB , and (Rh)lab discussed above, are:

1) Rydberg Constant for Infinite Mass Ro* : R« =nlc 3 me e*/Sh 3

where meis the mass of the electron. It is determined from

high precision laser spectroscopic measurements of the low-lying energy levels of the hydrogen atom and is presently known to an accuracy of 0.0010 ppm (one part in 109

). 2) Magnetic Moment of the Proton in H2 0 in Units of the

Bohr Magneton mp /mb : Mp /mb is determined by comparing the

electron spin-flip frequency in the hydrogen atom obtained us- ing a hydrogen maser operating at a magnetic field of several tenths tesla to the proton spin-flip frequency of protons in an H2 0 sample in the same magnetic field as measured by nuclear magnetic reasonance (NMR). Its current uncertainty is 0.012

ppm. (The Bohr magneton hb = eh/2nme .) 3) Molar Mass of the Proton Mp

: Mp

, the mass of one mole

of protons, is determined via mass spectrometry on the unified atomic mass scale defined such that the molar mass of the

12 C

atom is exactly 0.012 kg. It has an uncertainty of 0.012 ppm

at present.

4) Ratio of the Proton Mass to the Electron Mass mp /me

:

mp /meis obtained from measurements of the cyclotron fre- quencies of sequentially trapped protons and electrons in a high-magnetic field. Its present uncertainty is 0.043 ppm.

5) Inverse Fine-Structure Constant a' 1

: a

-1 = 2h/n0 ce

2 may be obtained by combining an experimental value for the anomalous magnetic moment of the electron ae determined

from measurements of various resonance frequencies of a single electron trapped in a magnetic field and the theoretical quantum electrodynamic or QED expression for a e

. The uncertainty of oT x (ae) is currently 0.073 ppm. A value of a"

1 with a current uncertainty of 0.16 ppm can be obtained by

combining measurements of the ground-state hyperfine splitting

of muonium ^Mhfs and the ratio of the magnetic moment of

the muon to that of the proton /i M

/jLip with the theoretical ex- pression for the splitting. (Muonium, Mu, is the ju +e~ atom

where ;u +

is the positive muon.)

6) Faraday Constant F: The Faraday constant in laboratory

units FLAB is determined electrochemically by measuring via a coulometer the number of moles of a substance such as silver deposited by a known charge. The number of moles is found

from the mass of the material deposited and its molar mass,

the latter determined in a separate experiment using mass spec- trometry. The charge is found from the current measured in terms of A LAB times its duration of flow. The most accurate determination of FLAB to date has an uncertainty of 1.3 ppm.

7) Gyromagnetic Ratio of the Proton in H2 O, yp . yp = co'p/B

where u>'p

is the NMR angular frequency of protons in H2 0 in an applied magnetic field B. It may be determined by two dif- ferent methods which in fact yield two different quantities

because of the way the laboratory or as-maintained electrical units enter the determination of B. 7p(low)LAB , where the "low" indicates a low magnetic field on the order of 0.001 T,

is obtained when B is calculated from the measured dimensions

of a precision solenoid carrying a current known in terms of

^lab- 7p(high)LAB , where "high" indicates a high magnetic

field on the order of 0.5 T, is obtained when B is determined

by measuring the force on a conductor of known dimensions

carrying a current known in terms of ^4lab- At present the most accurate determinations of 7p (low)LAB and 7p

(high) LAB

have uncertainties of 0.21 and 1.0 ppm, respectively.

8) Avogadro Constant NA : NA may be obtained from vari- ous measurements on a pure, highly perfect, single crystal. The

relevant equation for silicon is

[M(Si)/p(Si)]

VSd322Q (Si)'

(10)

Here Af(Si) is the molar mass of silicon, usually determined via mass spectrometry; p(Si) is its density, usually determined via hydrostatic weighing against standards whose densities have

been determined from their dimensions and mass; and d2 2o

is the (220) lattice plane spacing repeat distance determined

via combined X-ray and optical interferometry. The present

uncertainty inNA obtained this way is about 1.3 ppm.

Because the above electric-unit dependent constants (F and

7p) may not be measured in the same laboratory, they may not

be expressed in the same laboratory units. It is possible, however, to re-express all such quantities in terms of a single set of

17
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

laboratory units with little loss of accuracy by using the known

values of (2e/h) LAB and (Rh)lab 01 the results of direct unit comparisons carried out by means of transportable voltage and

resistance standards. In what follows it is assumed that all such

quantities have been so re-expressed.

Using the above fundamental constants one can derive a number of different equations for the quantities (A\0lab>

(Af^LAB, and (A^a)lab- For example for the volt one has

(*v)lAB

li 0 c

2 Mp (2e/h)lAB

1/2 00

J6R„(mp /m e )a 1 NA^ Based on the uncertainties given for the relevant constants entering this relation, it yields an indirect value of (A"v )lab with

an uncertainty of about 0.65 ppm, nearly four times less than

the uncertainty of the most accurate direct volt balance deter- mination. [In this equation and those that follow (2e/h)hAB

and (Rh)lab are taken as exact since they are adopted values used to define VhAB an(l ^lab- The resulting uncertainty for (Xv)lab, (Xn)LAB> or (A:a )lab is then the uncertainty in the relationship between the defined laboratory unit and the corresponding SI unit. If one wants the uncertainty in the

relationship between the laboratory unit as-maintained and

the corresponding SI unit then the uncertainty associated with

the implementation of the definition of the laboratory unit must also be taken into account by assigning an appropriate

uncertainty to (2e/h) LAB and (^)lab-j

For the ohm one may write

(*n)

Mo C(* LAB

2(RH) LAB

as well as

(*n)LAB

'nW{n'lnB ){2elh)LAB T/3

1 6R „ (RH)l AB j'p OOW)LA b

(12)

(13)

Equations (12) and (13) both yield an indirect value of

(Kq )lab with an uncertainty of about 0.07 ppm, comparable with the uncertainty of direct calculable capacitor

determinations.

Finally, for the ampere one has

(^a)lab

as well as

(^a)lab

7p(low)LAB

-| 1/2

_7p(high)LAB_

Mp7p(low)LAB (mp /me )(n'/iJiB )FhAB

1/2

(14)

(15)

where use has been made of the following relation to derive (15) from (14):

7n(high)LAB = F'LAB (n'/iJiB )(mp /me )/Mp (16)

(Since the uncertainties of /jlp/hb, rnp /me , and Mp

are all below 0.05 ppm, (16) implies that with little additional uncertainty a measurement of Flab may t> e viewed as a measure- ment of 7p

(high) and vice versa.) The uncertainty of the in- direct value of (A^a)lab fr°m 04) is about 0.5 ppm and that from (15) about 0.7 ppm. These uncertainties are significantly

less than the uncertainties of direct current balance determinations which are in the range 4-10 ppm.

A number of conclusions may be drawn from the above.

First, indirect realizations of the SI definitions of the volt and

ampere based on fundamental constant experiments are at present significantly more accurate than direct volt or current balance realizations. For the ohm the two different approaches

have comparable uncertainties. Second, the uncertainties associated with these indirect realizations are still rather larger

than the uncertainties associated with the implementation of

the definitions of J^lab and ^lab based on the Josephsonand

quantum Hall effects (perhaps a factor of 10 for the volt and

a factor of 2 for the ohm). Thus it is still not yet possible to achieve the desirable goal of having the as-maintained system

of electrical units consistent with the SI to the same level of

accuracy with which the as-maintained system can be implemented. Third, it is clear that the fundamental physical constants have a critical role to play in the maintenance and

realization of the electrical units and that it is a reciprocal rela- tionship-the accurate determination of fundamental constants

requires a reliable system of as-maintained units. Indeed, the

critical role the electrical units play in the fundamental constants field is clearly evident in the 1983 least squares adjustment of the fundamental constants as discussed in the following

section.

III. The 1983 Least Squares Adjustment

The purpose of a least squares adjustment of the fundamental physical constants is to generate a self consistent set of

"best values" of the constants for general use in science and

technology based on all the data available at a given epoch.

Comprehensive studies of the constants were pioneered in the

late 1920's by R. T. Birge and continued by others including

J. W. M. DuMond, E. R. Cohen, and Bearden and colleagues.

The most recent adjustment was carried out by Cohen and

Taylor in 1973 [5] under the auspices of the CODATA (Committee on Data for Science and Technology) Task Group on

Fundamental Constants. It served to update the comprehensive review and adjustment published by Taylor, Parker, and

Langenberg in 1969 [6] , and the set of best values resulting

from it was officially adopted for international use by CODATA

at its Eighth General Assembly in September 1973. This set is

still that currently recommended.

In the last decade or so an extraordinary amount of new

experimental and theoretical work relating to the fundamental

constants has been reported which now renders the 1973 set of recommended values obsolete. Thus the author, again in collaboration with E. R. Cohen and under the sponsorship and

guidance of the CODATA Task Group on Fundamental Constants, is undertaking a new adjustment to replace that of

1973. What follows is a brief summary of this effort (still not

quite complete at the time of writing) which will be called the "1983 Least Squares Adjustment of the Fundamental Physical Constants."

A. Some Definitions

A least squares adjustment of the constants is generally carried out by dividing the available data into two groups. One group

18
TAYLOR: ELECTRICAL UNITS, FUNDAMENTAL CONSTANTS

is known as the "stochastic input data" and consists of the less precisely measured quantities entering the adjustment and

which are subject to adjustment. Examples from the quantities

already discussed include (Kn ) LAB , FL AB ; [7p(high)] LAB , and

In the 1983 adjustment the electric-unit dependent stochastic data are expressed in BIPM units as they existed on January 1

,

1983, namely, FBI83 , ft BI83 , and^ BI83 = VBl83 /nBl83 . The

BIPM has been maintaining its laboratory unit of voltage via the Josephson effect since January 1 , 1976, with the adopted value (2e//07 6-Bi = 483 594 GHz/F76 . BI . Defining Kv = VB183 /V

it then follows that

*V = VBl83 IW

= V16 . m/V=(2e/h)16 . Bl l(2e!h) = 483 594 GHz • Vi6\ Bl /(2e/h).

(17a)

(17b)

(17c)

The BIPM laboratory unit of resistance is currently defined in terms of the mean resistance of six l-ft resistors so that fi Bi 83

is based on the mean value of these resistors on January 1, 1983. Thus one simply has

— ^BI83/^

and

^A = ^ BI83/A = Ky/Kn .

(18)

(19a)

(19b)

The second group of data is termed the "auxiliary constants" and consists of quantities which are sufficiently accurate in comparison with the stochastic data that if included in the adjustment as stochastic data, their adjusted (or output) values and uncertainties would change negligibly compared with their input values and uncertainties. Thus the auxiliary constants

are not subject to adjustment. Examples from the quantities

already discussed include \ip

\\iB ,Mp

, and mp /me

. The two main selection criteria for considering individual items for inclusion in the 1983 adjustment are (1) availability

prior to December 31, 1983 due to publication deadlines and

the time required for review and analysis; and (2) the requirement that the item make a nontrivial contribution to the ad- justment. For example, if a direct measurement of a quantity

contributes only a few percent to the determination of the ad- justed value of the quantity, then it should be excluded because

it is clearly not a competitive determination. (In a least squares adjustment w(

- = 1/s? where w,- is the weight of the ith stochastic input datum and st its one standard deviation uncertainty.)

B. The Data and Their Consistency

At the time of writing 12 different types of stochastic input

data are being considered for inclusion in the 1983 adjustment

with a total of 34 individual items. Some older measurements

of comparatively large uncertainty are initially being included

in the hope that they might clarify a number of discrepancies

which exist among the more accurate data. These 34 items are

summarized in Table I. The two values of a

-1 listed are a-1 (ae ) and a

_1

(4 He) as obtained from the experimental determination of, and theoretical expression for, the anomalous magnetic

moment of the electron and a certain fine-structure splitting in atomic helium, respectively.

A detailed discussion of the consistency of these data is be- yond the scope of this paper. However, a few highlights will be

given in order to further indicate the many inter-relationships

which exist among the constants and the critical role of the

electrical units and electric-unit dependent quantities. First we comment on the consistency of like data, for example, the

six values of 7p (low) BI83 , and then on the consistency of unlike data as implied by various expressions involving different constants, such as (1 1) and (16) and others to be given below.

On the assumption that only a difference between two quantities two or more times their combined standard deviation uncertainty sc = [s\ + s\] 1/2

is statistically significant, the main

like-data inconsistencies are between the several values of y'p (both low and high) and the two values of c? 2 2o(Si)- For ex- ample, the most accurate value of 7p(low) BI83 differs from the two next most accurate values by I2sc and 7.6sc , respectively,

and these two differ from each other by 2.4sc ; the first and

second most accurate values of 7p

(high) BI83 differ by 2.4sc ; and the two values of d22 o(Si) differ by 7.7sc

. The unlike data may be compared in many different ways;

the focus here will be on those which tend to clarify the in- consistencies between like data. We first note that the indirect value of 7p

(high) Bi 83 obtained from FBI83 via (16) is in excel- lent agreement with the most accurate of the four direct measurements; the two differ by only 0.8sc

. Thus the second most

accurate value of 7p(high) is suspect (see previous paragraph).

Next we compare the two direct values of a

-1 listed in Table I

,

namely, a~

l (ae ) and a

-1

(4 He), with 10 indirect values derived

from the following expressions using the data of Table I:

a

l (RH) = 2(RH) Bls3 Ka lnQ c

a

_1 (7P ) =

c(jU ;/juB )(2e//2) 76 . BI-I

1 / 2 4#»tfn 7p(low)BI 83

a

-1 (Mu) = [CMhfs (MM

/Atp)/tfMhfs ] 1/2

(20)

(21)

(22)

where CMhfs I s obtained from the theory of the muonium

hyperfine splitting and the ratio CMhfs / uMhfs has an uncertainty

of 0.14 ppm. To obtain the three quantum Hall resistance and

six proton gyromagnetic ratio values of a"

1 from (20) and (21),

respectively, the weighted mean of the five individual, relatively consistent values of is used (uncertainty < 0.07 ppm). To

obtain the single muonium hyperfine splitting value of a

-1 from

(22), the weighted mean of the two individual and consistent

values of /uM

/jUp is used (0.3-ppm uncertainty). The result of

the comparison is that all 12 values of a

-1 are reasonably consistent except the two obtained from the two next most ac- curate values of 7p(low)BI83 . Thus these values of 7P (low)BI83

are definitely suspect. It is significant that the uncertainty of

a number of these indirect values of a

-1

is comparable with

the 0.073 ppm uncertainty of a

-1 (ae) (within a factor of 2-3)

and a value with essentially the same uncertainty as <x~

1 (a e )

19
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

TABLE I Summary of Tentative Stochastic Input Data for the 1983 Least Squares Adjustment

Q u3

n

X. i ty Mniii- 1 u

ri f W 1 1 113 C i valued U ML tr 1 La 1 1 1 L 1 CD (ppm)

Ko5 0.11; 0.12; 0.18; 0.22; 0.36

KA

6 4.0; 5.8; 6.1; 7.7; 7.9; 9.7

K v

1 2.4

F BI83

1 1.3

^p (low) BI83 6 0.24; 0.52; 0.61; 0.82; 2.1; 3.2

Y p (high) BI83 4 1.0; 3.2; 3.6; 7.5

d 22Q(Si) 2 0.1; 0.21

M(Si)/p(Si) 1 1.15

(,VbI83 3 0.11; 0.14; 0.32

-1

a 2 0.073; 0.78

M P

2 0.36; 0.53

VMhfs 1 0.036

and also consistent with it can be derived from

a

l (RH/yp ) =

'(^d/Mb) (Rh)bi 83(2e//2) 76 -Bl' 1/3 (23)

2/Jt 0 /? Oo7p(low)BI8 3

using the most accurate value of (Rh)bis3 and 7p0ow)Bi83-

We next compare the unlike data via the values of NA they

imply. Seven such values may be readily obtained from the data of Table I, two from (10) using the two different values of <i 2 2o(Si) and the single value of Af(Si)/p(Si); one from

NA =

Moc(2e//2) 76 . BI a

-1 FBi 83 Kn

and four from

_jHocMp(2e/ft)7 6-Bitt~'7p(high)Bi83

(.v'P l»B)(mplTMe)Kn

(24)

(25)

where in both these equations the weighted mean value of

is used and a' l (a e) is taken for a

-1

. The result of the comparison is that the three values of NA obtained from FBi 83 , the most accurate value of 7p (high)Bi 83 , and one of the two

values of <i 2 2o(Si) are in good agreement (and have uncertainties between 1 .0 and 1 .4 ppm as well), but disagree with the value of NA obtained from the other measurement of c? 220 (Si). The

latter is thus suspect and the second most accurate value of

7p (high) BI83 is definitely questionable.

Finally, we compare the unlike data via the values of Kv

they imply. Fourteen values may be readily derived from the

data of Table I which together with the one direct measure- ment gives a total of 15. Six values follow from the direct measurements of KA via Kv = K\ Kq . (26)

Four values from measurements of 7p(low) Bi 83 and7p

(high) Big3 carried out in the same laboratory using the relation

Kv = Kn7p(low)BI83

'~1 1/2

7p(Wgh)BI83 (27)

[see (14) and (26)] ; one value using (27) with 7p

(high) BI83 re- placed by FBi83 via (16) and where both 7p (low)Bi 83 and

FB i 83 were measured in the same laboratory; two values using

(11) with NA obtained from (10) using the single value of

M(Si)/p(Si) and the two values of tf2 2o(Si); and one value from

Kv = Mp c(2e/h)16 . mKn

Hi/a

4R 00 (mp /me )(a- l YFBl83 (28)

where here and in the other equations (as appropriate) the weighted mean is used for and a

-1 (a e ) for a-1

. The result of this comparison tends to confirm most of what had been

concluded from the earlier comparisons, namely, that the two

next most accurate values of 7p (low)Bj 83 are inconsistent with

the most accurate value and that the most accurate value for 7p

(high) B183 , the value of FBi 83 , and one of the values of

<^2 2o(Si) are highly consistent but that the other value of

20
TAYLOR: ELECTRICAL UNITS, FUNDAMENTAL CONSTANTS

^22o(Si) is in considerable disagreement with these data. Unfortunately, however, the values of Ky derived from the direct measurements of KA via (26), as well as the single direct measurement of Ky, have such large uncertainties that most are consistent with the majority of the indirect values. Thus at the present time they are not capable of clarifying (i.e., either supporting or refuting) the 7p(low)Bi 8 3. 7P

(high)BI83 , and

^22o(Si) inconsistencies.

C. Least Squares Adjustment

To carry out a least squares adjustment of the constants,

a subset of constants, known as the adjustable constants

or "unknowns," is chosen in terms of which stochastic in- put datum can be individually expressed, if necessary with

the aid of the auxiliary constants. The adjustable constants

are selected in such a way that the resulting equations, known

as the observational equations, are statistically independent.

The total number of observational equations is thus equal to the number of items of stochastic data. Substituting the numerical values for the auxiliary constants and stochastic data and using a computer, the observational equations may be

solved for the unknowns and their uncertainties. (The least squares procedure automatically takes into account all possible "routes" to the adjustable constants, both direct and indirect.) In the 1983 adjustment the adjustable constants or unknowns

are most likely to be taken as of 1

, K\, Kn , c? 2 2o(Si), and

M/i/Mp- The resulting 12 different observational equations

(34 total) for the 12 different kinds of stochastic data listed in Table I would then be as given in Table II. The least squares adjustment using all 34 data items of

Table I has x

2 of about 320 for 29 degrees of freedom. The

gross disagreement of some of the data discussed is, of course,

responsible for such a large (statistically improbable) x

2

• In the last several years much work has been done on the problem of how best to handle inconsistent data in a least squares adjustment of the constants [7] . While final decisions have not yet been made regarding the 1983 adjustment data,

based on the analyses done to date it may tentatively be con- cluded that the 1983 recommended values for many constants

will differ significantly from their 1973 counterparts. For ex- ample, it is likely that e will decrease by approximately 7 ppm

and that its uncertainty will decrease from nearly 3 to less than 0.5 ppm; for h the comparable changes will be -15 ppm

and from 5.4 to less than 1 ppm; and for NA

+ 15 ppm and

from 5.1 to less than 1 ppm.

These changes come about because e, h, and NA are related to the adjustable constants a

-1 and Ky through

e = [(Moc/4) (2 e/h) 16 . Bl ] ~l (a' 1 )"» Kw (29)

h = [fi Q c/2] a"

1 e

2 (30)

NA = [Mp l(mp lme)noR~] (a" 1 )" 3 ^

2

. (31)

The values of the quantities in brackets and a" 1 will probably

change little compared with their 1973 values-tenths of ppm

at most (with the exception of mp /me , all the quantities in brackets were auxiliary constants in the 1973 adjustment as well as in the 1983 adjustment). However, it is likely that the adjusted value of Ay will decrease by approximately 7.5 ppm

TABLE II Probable Observation Equations for the 1983 Least Squares

Adjustment"

Kv

Kn

(o

" 1)

" 2kv\

{K

A

}

4R (m /m )

°° P e

j c . M

p c(2e/h)

76-BI 8183

4R

c( Mp /M B )(2e/h)

76 . BI

4R

{ *p (1 °W) BI83 !

c(M>B )(2e/h)

76 . BI fVhi 9

h >BI83*

d22Q(Si) = {d220

(Si) }

( a~V\2 d

320

(Si)

16R (m /m ) Voc2(2e/h)

76-BI

M(Si)

-!

} p(Si)

-1

U /M_

M P

(a _1 )- 2 (p

M

/Mp ) = C^fs

{vMhfs }

The curly brackets indicate numerical value only.

and its uncertainty from 2.6 to less than 0.5 ppm due to the new and highly accurate indirect values of Ay implied by (1 1), (27), and (28). From (29), (30), and (31) it follows that a

7.5 ppm decrease in the value of Ay will lead to a decrease in the value of e of the same amount, twice as large a decrease in

h, and twice as large an increase in NA . Moreover, since the 1983 adjusted value of a

-1 will probably have an uncertainty

of less than 0.1 ppm, the uncertainty in e will be about the same as that in Av and the uncertainties in h and NA will be

about twice as large.

It is important to realize that the approximate 7.5 ppm change

in Ay likely to result from the 1983 adjustment implies that the value of 2e/h in general use for volt maintenance pur- poses using the ac Josephson effect is smaller than the SI value by this amount. As a consequence, the as-maintained units of

voltage of the various national laboratories are smaller than

the SI volt by about 7.5 ppm. A readjustment of the various laboratory units to bring them into better agreement with the

SI system is expected within the next few years [8]

.

IV. Future Electrical Measurements Research

We conclude with a brief discussion of future electrical measurements research which can improve our knowledge of the fundamental physical constants. Of highest priority would seem to be the direct measurement,

21
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

via a force balance experiment or its equivalent, of either (Xv)lab or (Xa)lab t0 an accuracy of 0.1-0.3 ppm. Such

a measurement could unequivocally resolve the present dis- crepancy between the two available values of d22 o(Si) [see (10), (1 1), (26)] as well as clarify the discrepancies associated with the two next most accurate values of 7p(low) [see (14),

(27)]

.

A new direct determination of 7p(low)LAB to an accuracy

of 0.1 ppm could also clarify the current 7p(low) problem,

while if coupled with a new measurement of either 7p

(high)LAB

or ^lab with 311 accuracy of 0.5 ppm, could contribute to the

resolution of the rf 2 2o(Si) problem [see (10), (1 1), (14), (15),

(26)]

.

Measurements of (Rh)lab and (Xn)LAB with uncertainties

at the few hundredths ppm level could also be of major importance. For example, in combination with the new measure- ment of 7p(1ow)lab advocated above such a value of(i?//) LAB

would provide the means for determining a

-1 to an accuracy

considerably less than 0.05 ppm and one which does not re- quire the absolute realization of an electrical unit [see (23)]

.

This value could then be compared with that obtained directly

from (RH) LAB and (Kn ) LAB [see (20)] as well as a.~

l (ae ) (0.073 ppm current uncertainty and soon to be reduced by

about a factor of 3). Such over-determination or "redundancy"

is of critical importance in the fundamental constants field since agreement between values obtained by a variety of methods

is the only way one can ensure that there are no unsuspected

systematic errors plaguing particular experiments.

Finally, work needs to be done to develop improved trans- port standards, especially for resistance, so that the results of

absolute ohm realizations and quantum Hall resistance measure- ments carried out in different laboratories can be critically

compared. Since it is unlikely that transport standards with

the requisite parts in 10 8 stability upon shipment can be de- veloped as readily for voltage as for resistance, new direct comparisons of the Josephson volt standard apparatus of the

various national laboratories should be undertaken to ensure

that the claimed accuracies of a few parts in 108 are justified.

Such comparisons of HlAB and FLAB could contribute signifi- cantly to improving our confidence in the consistency of the laboratory electrical measurement systems of the various countries throughout the world, as well as in the 1983 and

subsequent sets of recommended values of the fundamental

physical constants.

Note Added in Proof. Since this paper was submitted for publication, it was decided to consider data available through

early 1985 for possible inclusion in the new adjustment and

to rename it the "1985 Least Squares Adjustment of the Fundamental Physical Constants." However, the paper's basic contents and conclusions are unaltered by this decision.

References

[1] The International System of Units (SI), P. T. Goldman and R. J. Bell, Eds., Natl. Bur. Stand. (US), Spec. Publ. 330. Washington, DC: U.S. Government Printing Office, 1981.

[2] "Documents concerning the new definition of the meter," Metrologia, vol. 19, no. 4, pp. 163-177, 1983.

[3] It is not the purpose of this tutorial to provide a comprehensive

bibliography. Rather, the reader is referred to [ 5] and [6] , and to various relevant articles in this volume and in the following ref- erences: Atomic Masses and Fundamental Constants 5, J. H. Sanders and

A. H. Wapstra, Eds. New York: Plenum, 1976. Metrology and Fundamental Constants, Proc. of the International School of Physics "Enrico Fermi," Course LXVIII, A. Ferro Milone,

P. Giacomo, and S. Leschiutta, Eds. Amsterdam: North Holland, 1980. See especially, the article by B. W. Petley, "Electrical metrology and fundamental constants," pp. 358-463. Atomic Masses and Fundamental Constants 6, J. A. Nolen, Jr. and W. Benenson, Eds. New York: Plenum, 1980.

Precision Measurement and Fundamental Constants II, B. N. Taylor and W. D. Phillips, Eds., Natl. Bur. Stand. (US), Spec. Publ. 617. Washington, DC: U.S. Government Printing Office, 1984. Quantum Metrology and Fundamental Physical Constants, P. H.

Cutler and A. A. Lucas, Eds., NATO Advanced Study Institute

Series, Series B: Physics, vol. 98. New York: Plenum, 1983. The Fundamental Physical Constants and the Frontier of Measurement, B. W. Petley. Bristol: Adam Hilger, 1984.

[4] K. v. Klitzing, G. Dorda, and M. Pepper, "New method for high ac- curacy determination of the fine-structure constant based on quantized Hall resistance," Phys. Rev. Lett., vol. 45, no. 6, pp. 494-

497, Aug. 11, 1980. See also the several papers on the quantum

Hall effect in this volume.

[5] E. R. Cohen and B. N. Taylor, "The 1973 least-squares adjustment of the fundamental constants," J. Phys. Chem. Ref. Data, vol. 2, no. 4, pp. 667-734, 1973.

[6] B. N. Taylor, W. H. Parker, and D. N. Langenberg, "Determination

of e/h, using macroscopic quantum phase coherence in superconductors: Implications for quantum electrodynamics and the funda- mental physical constants," Rev. Mod. Phys., vol. 41, no. 3, pp. 375-496, July 1969. Also published as The Fundamental Constantsand Quantum Electrodynamics. New York: Academic, 1969.

[7] B. N. Taylor, "Numerical comparisons of several algorithms for treating inconsistent data in a least-squares adjustment of the fun- damental constants," Natl. Bur. Stand. (US), Report No. NBSIR

81-2426, Jan. 1982.

[8] P. Giacomo, "News from the BIPM," Metrologia, vol. 20, no. 1, pp. 25-30, 1984.

22
NBSIR 75-935

THE NATIONAL MEASUREMENT SYSTEM

FOR ELECTRICITY

Norman B. Belecki

Bernadine L. Dunfee

Oskars Petersons

Electricity Division

Institute for Basic Standards

National Bureau of Standards

Washington, D C. 20234

Issued September 1978

U.S. DEPARTMENT OF COMMERCE, Juanita M. Kreps, Secretary

Dr. Sidney Harman, Under Secretary

Jordan J. Baruch. Assistant Secretary for Science and Technology

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director

23
PREFACE

The National Measurement System for Electricity is that system of people, organizations,

standards, apparatus, and software which performs the function of providing quantitative information on materials, manufactured items, and processes through the measurement of their

physical properties or parameters. This report is one of a series of similar reports pro- duced by the Institute for Basic Standards (IBS) of the National Bureau of Standards to provide a complete description of that system, of which IBS is the foundation. The format of

this report is common to the series and was chosen to provide uniformity and completeness.

The authors would like to express their appreciation to their many colleagues who supplied information and perceptions for inclusion in this study. Special appreciation goes to those who arranged or hosted visits at private and government organizations for the authors

and others of the Division staff. We would like to also acknowledge the work of Denise

Prather, who assisted in the production of the final draft. Finally we are most indebted

for the efforts of Sylvia Ramboz, who was responsible for the production of the manuscripts

for two interim reports as well as this document; and for her comments, suggestions, and

encouragement.

24
CONTENTS

PREFACE

EXECUTIVE SUMMARY

1. INTRODUCTION

2. STRUCTURE OF THE ELECTRICAL MEASUREMENT SYSTEM

2.1 Conceptual System . , 2.2 Basic Technical Infrastructure . 2.2.1 Documentary Specification System

2.2.1.1 Standardization Institutions , . . . 2.2.1.2 Survey of Documentary Standards

2.2.2 Instrumentation System

2.2.2.1 Measurement Tools & Techniques

2.2.2.2 The Instrumentation Industry . . 2.2.3 Reference Data 2.2.4 Reference Material . . 2.2.5 Science and People

2.3 Realized Measurement Capabilities ...

2.4 Dissemination and Enforcement Network

2.4.1 Central Standards Authorities

2.4.2 State and Local Offices of Weights

and Measures

2.4.3 Standards & Testing Laboratories & Services . . . . 2.4.4 Regulatory Agencies . 2.5 Direct Measurements Transactions Matrix . . . . . 2.5.1 Analysis of Suppliers and Users 2.5.2 Highlights re Major Users

3. IMPACT, STATUS AND TRENDS OF the MEASUREMENT SYSTEM

3.1.1 Functional, Technological, and Scientific

Applications

3.1.2 Economic Impact - Costs and Benefits

3.1.3 Societal and Human Impacts

3.2 Status and Trends of the System

4. SURVEY OF NBS SERVICES

4.1 The Past

4.2 The Present Scope of NBS Services

4.2.1 Description of NBS Services

4.2.2 Users of NBS Services

4.2.3 Alternate Sources . . . 4.2.4 Funding Sources for NBS Services

4.2.5 Mechanisms for Supplying Services .... . 4.3 Impact of NBS Services

4.3.1 Economic Impact of Major User Classes . . .........

4.3.2 Technological Impact of Services ......

4.3.3 Pay-off from Changes in NBS Services

4.4 Evaluation of NBS Programs

4.5 The Future . .

5. SUMMARY AND CONCLUSIONS

25
APPENDIX A - The Measurement System Study

APPENDIX B - NBS Primary and Working Electrical Standards . . . .

APPENDIX C - "Critical Electrical Measurement Needs and Standards

for Modern Electronic Instrumentation" - Workshop

Report Abstract, Forward, and List of Attendees . .

APPENDIX D - Standards and Calibration Laboratories

APPENDIX E - Typical Measurement Support Structures

REFERENCES

BIBLIOGRAPHY

LIST OF FIGURES

Figure 1. The derivation of the electrical units

Figure 2. Traceability of voltage measurements . Figure 3. Traceability of energy measurements . . Figure 4a. Uncertainties of measurements

Figure 4b. Uncertainties of measurements

Figure 5. Typical corporate support structure . . Figure 6. Organizational transactions matrix . . Figure 7. Code for transactions matrix

LIST OF TABLES

Table lo Electrical quantities and units . . . . . . . Table 2 Characteristics of the basic legal units . . . Table 3. IEC committees affecting the

Electrical Measurement System . . . „ . . . . Table 4. OIML Pilot Secretariat 13 . „ ....... . Table 5. Voluntary standards sponsoring organizations , Table 6. Basic units and industrial standards . . . . . Table 7. List of common instruments ..........

Table 8. List of electrical standards manufacturers „ . Table 9. Sample sources of reference data for engineers

Table 10. Regulatory agencies and agency categories „ . Table 11. Representative heavily measurement-dependent

products ...................

Table 12. Electricity Division technical outputs . . . . Table 13. Dissemination services of the

Electricity Division ...... . Table 14. Users of NBS Services ............

Table 15. Electricity Division user classes ......

26
THE NATIONAL MEASUREMENT SYSTEM FOR ELECTRICITY

Norman B. Belecki

Bernadine L. Dunfee

Cskars Petersons

Electricity Division

Institute for Basic Standards

June 1976

EXECUTIVE SUMMARY

Electrical measurements are of critical

importance due to the universal use of electricity as the primary means for the trans- mission and control of energy and data. The

National Measurement System for Electricity

is that set of laboratories, organizations,

documents and people which make possible

and are responsible for all electrical mea- surements made in the U.S. at all levels of

accuracy. This document summarizes the re- sults of a four-year study of that system.

The structure of the system may be con- ceived as having a number of components,

each serving a particular sector of society

and each having distinct characteristics re- flecting differing requirements. Two major

components exist: the industrial electronics

component and the electric power industry

component. There are a number of minor

parts in each.

Electrical measurements in the system's

industrial electronics component are used to ensure quality and reliability of materials

and manufactured products, to provide for

interchangeabil ity of components and parts,

and to control meanufacturing processes.

This component can also be further subdivided into two general areas: the measure- ment-intensive industry segment and the general industry segment. Measurement-intensive industry and its measurement system are characterized by high technology and the need

for a very high degree of product reliability.

Included in this portion of the defense, com- munication, computer, and electronic component

industries. Measurement accuracy within this

segment is ensured by a hierarchical laboratory system in each organization. Test equipment used for quality and process control is calibrated periodically using precision instruments which are in turn calibrated by

comparison with primary standards in the cor- porate standards are then periodically calibrated at the National Bureau of Standards or

i ntercompared regularly with standards which

have been so calibrated. This system, which

in large organizations can have a complex

hierarchy of laboratories, is primarily the

result of the quality control requirements

of DoD, RSA, and NASA procurement contracts,

the reauirements of regulatory agencies such

as EPA and OSHA and logistics constraints.

In the general industry segment of the

system, auality control requirements are usually less stringent than in the high

technology segment; accuracies are generally

lower; and the manufacturing processes are

less affected by absolute accuracy of mea- surements. Thus, organizations in this

portion of the system are inclined to buy

calibration and repair services from instrumentation manufacturers, large corporate

laboratories in the aerospace industry, or companies specializing in calibration and

repair work. As previously noted, each of

these generally maintains a rigid, well-defined measurement support system.

There appears to be a number of problems

and shortcomings in the industrial electronics component of the system. Among the most

important are a lack of standards (written

and artifact) for the support of dynamic,

high-speed electrical measuring instruments

and automated test systems; the non-availability in certain areas of procedures and

test techniaues of guaranteed reliability;

a potential future shortage of competent

measurement personnel; a failure on the part

of some people in the system to perceive

that calibration is not always a sufficient

condition for assurance of measurement auality; and an inherent rigidity in the government contractural requirements which stifles

innovative approaches to measurement prob1 ems.The electric power industry segment of

the National Measurement System for Electricity can also be divided into two parts:

operational and research for improved electric power transmission and distribution.

Electrical measurements play an important

role in the day-to-day operation of the na- tion's electric power systems. They are

used to control the generation and trans- mission of electricity, to provide a basis

for equitable exchange (energy metering),

and for testing machinery and equipment sup- plied to the power companies.

27
Measurement support for both of these areas is generally provided by the measurement laboratories of power companies and

those of the electrical equipment manufacturers. In the energy metering area, many

of the state and local public utility com- missions which regulate the power companies

require acceptance testing of watthour meters as well as retesting of older meters.

NBS calibration of physical standards and

electrical apparatus, both at NBS and in the

power companies' and manufacturers' laboratories, ensures the overall integrity of the

system.

Most of the industry - both utilities and

manufacturers - is adequately equipped to measure traditional quantities such as current and voltage at levels up to 15 kilovolts. Fewer companies, especially in the

utility sector, have capabilities above 100

kilovolts. At extra-high voltage (EHV) and

ultra-high voltage (UHV), the utility companies, with rare exceptions, have little

measurement and calibration capability.

Such capability, however, does exist with

the manufacturers of large electrical equipment. With respect to nontradi tional quantities, such as transient high voltages and

currents, measurement capability is generally

available only with the large equipment manufacturers.

The quality and consistency of electric

power system-related measurements, including

traceability to NBS-maintained standards,

follow the above pattern of measurement ca- pability. Whenever the calibrations are relatively simple and inexpensive, and if

there is also regulatory incentive to do

adequate work, the system is in excellent

condition. A good example is watthour meter

calibrations. But for higher voltage, and

especially for non-conventional measurements

where the calibrations are difficult to perform and reliable methods are not always

readily available, the system has deficiences. Included in this category and EHV and UHV steady-state measurements, and transient measurements in general.

Electricity will become the predominant

form of energy in use before the end of the

century. Consequently, a large research and

development effort funded by the Energy Re- search and Development Administration and

the Electric Power Research Institute is being directed at ways to increase the transmission capability and efficiency of the

country l s electric power systems. New high

voltage and electrical -related measurements

are required as transmission technology advances. Examples of new technologies include cryogenic transmission systems, UHV overhead lines, compressed-gas insulated

substations, and high-voltage directcurrent transmission systems. Each is a relatively measurement-intensive, emergingtechnology area. While some older types of

measurement methodologies can be adapted

for these new areas, generally they cannot

and some significant deficiencies now exist including measurements for accelerated

aging tests of electrical insulation, and

traceability to national standards for impulse measurements.

Two important additional components of

the National Measurement System for Electricity are associated with the scientific

community and consumer electronics. Measurements made in laboratories not under DoP or NASA contract are generally supported by

local instrumentation shops. In contrast

with the usual practice in the industrial

electronics component, periodic recalibration is unusual - calibration is generally

performed only before and after a crucial

experiment, or as part of a special maintenance effort. Measurements in support of consumer electronics, such as TV, stereo systems, citizen band and amateur radio, auto

electronics, and general electrical work are usually of low accuracy. Consequently, manufacturers' claims of test equipment accuracy

are accepted until malfunction occurs. This

portion of the system has few, if any, serious measurement Droblems.

In summary, the Electricity Division of

NBS provides the basis for that uniform system of electrical measurement in the U.S. which permits equity in trade, interchangeability of components, the transmission and

distribution of electric power, a means of

maintenance for electronic equipment, the

ability to control the quality of production, and the general advance of science

and technology. This is achieved by developing improved means for realizing, maintaining, and disseminating the basic electrical

units; by calibrating electrical standards,

instruments, and systems for industry, government, and the academic community; by

developing new instrumentation and measurement methods; by obtaining basic data and

determining fundamental constants related

to the electrical units; by ensuring that

electrical measurements carried out in the

U.S. are consistent with those made in other

countries; and by providing advisory and con- sultative services.

28
1. INTRODUCTION

The National Electrical Measurement System began to assume its present form around"

the beginning of this century. At that time

the uses of electricity were limited to the

easy transmission and use of energy in manu- facturing and transportation and for lighting and primitive means of communication

(telephone and telegraph). The establishment in 1901 of the National Bureau of Standards led to the foundation of a uniform

system of electrical measurement where none

had existed before; a foundation which permitted and expedited the development of

electrotechnology which has made our present

way of life possible. Since then the mea- surement system has continuously grown in magnitude and scope along with the industries

it supports [1].

Today the system pervades every aspect of

modern life. The electric power industry is the largest in the United States [2]. The

electronics industry is one of the larger

industries as well, and its products make

possible the inexpensive production of goods

and services throughout the rest of industry. Our present level of electrotechnology

is only possible because a uniform system of

electrical measurement exists. This system

provides scientists, electrical engineers,

and quality control engineers with a common language and thereby minimizes the need for

trial-and-error approaches to design by permitting the existance of necessary data

bases and allowing the meaningful communication of technical information in quantified

detail. It is also true that the level of

sophistication and general quality of all electrical-electronic products as well as many others are predicated, among other factors, on the state of the measurement system. The stability with time of the units

of measure, the compatibility of measurements throughout the system, and the effectiveness with which new requirements are anticipated and dealt with are key factors

which establish bounds on the level and rate of advance of technology. The system per- mits interchangeability, at affordable cost,

of parts with very high levels of precision,

a vital part of modern manufacturing technology.

The study of the National Electrical Measurement System, the results of which are summarized in this report, is intended to provide a complete picture of the system at the present time, and to forecast its future

posture, needs, and shortcomings, The present intensive study is also to provide a data base for program planning within the

National Bureau of Standards, a data base

which will be expanded over the years as the

result of a continuous surveillance of the

system. The study was carried out primarily

in two parts. Technical data were gathered

as the result of visits to companies, corres- pondence with colleagues, interviews with

visitors to the NBS, and interactions with

fellow members of standards committees by

Electricity Division personnel. Economic

data were gleaned primarily from the literature. Because of the nature of the Electrical Measurement System, separate studies were not conducted on each measurement area. Rather, the system was observed as consisting of two uniaue sets of users, the electric power industry and the electronics industry. The term "electronics industry" as used here refers not only to the manufacturers of electronic components, modules,

instruments, and systems, but to the users

of these as well. Each of these will be

further subdivided as the report progresses.

Since the system is so large and pervasive,

no attempt was made to define its size ex- actly. Recurrina structural features make

sampling an effective means of viewing the

system.

As a result of the study, it was determined that:

(a) the system is adequately equipped and

manned to support most measurements of a static nature;

(b) although most new instruments are designed and specified for use under computer

control, there is virtually no capability in the system as a whole to support dynamic measurements; i.e., measurements in which time

is a factor;

(c) there are expected to be increasing

pressures from government regulatory agencies and from industrial firms engaged in international commerce for the establishment

of a system for accrediting standards, calibration, and testing laboratories of various

types;

(d) future sophisticated instrumentation,

if not designed for computer control, will

contain small control and loqic units called

microprocessore , creating a need for methods

of verification of measurement implementation software;

(e) there exists a need, whose urgency is being determined, for improved measurements

of electrical conductivity for application

to non-destructive materials testing;

(f) because of the energy crisis, there

is a need for new measurement methodologies

for the support of energy-related research,

especially in the areas of electric power;

(g) there is a lack of capability within

the electric power industry to perform ultrahigh voltage measurements;

(h) the exchanoe of technical information

takes place in the system at an unacceptably

low level.

29
It must be noted that consideration in this study was not given to those electrical

measurements associated with the processing

of semiconductor materials. Such measurements are addressed in depth by the impact

study of the Electronic Technology Division

of NBS.

2. STRUCTURE CF THE ELECTRICAL

MEASUREMENT SYSTEM

2.1 Conceptual System

The system provides the capability for measurement of electrical quantities shown in table 1 below in the frequency range between

dc and 1 MHz.

Table 1. Electrical quantities and units

Quantjty_ Uni ts

Capacitance (C) Farad

Current (I) Ampere

Charge (Q) Coulomb

Potential Difference (E) Volt

Inductance (L) Henry

AC-DC difference, voltage (Unitless

)

AC-DC difference, current (Uni tl ess

)

AC-DC difference, resistance (Unitless)

Power Watt

Energy Watthour

or Joule

Ratio, resistive (Unitless)

Ratio, dc voltage (Unitless

)

Ratio, ac voltage (Unitless

Ratio, current (Unitless

)

The above are measured in terms of the so- called legal units, those embodied in physical standards maintained by the National

Bureau of Standards or defined by them as a function of physical constants. These units are - to the degree practical - compatible

with the SI (Systeme International) units.

The Systeme International de'Unites provides the internationally accepted theoretical basis for the metric units of measure.

In it, seven base units - the metre, the

kilogram, the second, the kelvin, the mole,

the candela, and the ampere - are regarded

as being dimensionally independent and all other units stem from them, either through

known physical relationships or defining

equations [1], In practice a set of four

independent equations is available to define

the six basic electrical quantities given at

the top of table 1 with their symbols. They

are E= IR (Ohm's Law); Q= I t (t is time);

C=Q/E; and L=E/-^. These can serve to define only four of these auantities. The

system is completed by requiring the unit of

electrical energy to be equal to the unit of

mechanical energy and devising electromechanical experiments - absolute experiments - involving force and lenqth (whose product is energy) and corresponding electrical quantities to realize in practice two of the

electrical units.

Absolute experiments are very complex and

time-consuming, require the most meticulous

approach, and involve the design and con- struction of special apparatus. Moreover,

in the case of the ampere and volt, the un- certainties of the results are very large by

comparison with those attainable using

either artifacts, such as standard resistors

or capacitors, or physical phenomena (e.g.,

the Josephson effect) to carry forth defined

units. It is for these reasons our system

of legal units exists. ' Figure 1 shows in detail the derivation of assigned values for

the legal electrical units. In the past,

absolute experiments have been performed to realize the ampere by comparing the force

between current-carrying coils of known geometry to that exerted by the earth's gravitational field on a known mass. This was done most recently at NBS in 1 9F7 using a Pellat-type electrodynomometer. The result

was the determination that the ratio of the

NBS ampere (derived from the legal units) to the SI ampere equals 1.^000105 with a probable error of 7 parts in a million.

These results as well as those of four

other experiments formed the basis for an eleven part per million adjustment in 1969

to the value of the ampere as derived from

the units maintained in Sevres by the Bureau

International des Poids et Mesures - the In- ternational Bureau of Weights and Measures,

which by the treaty of the Metre maintains

electrical and physical units to promote international measurement uniformity. In con- sequence of this adjustment and drifts in the U.S. volt measured with respect to the BIPM volt, the U.S. leqal volt was changed

by 8.4 parts in a million on January 1, 1969.

U.S. laboratories were informed of this

change and advised to make the appropriate

adjustments to the assigned values of their

standard cell emfs through correspondence

and technical publications in the media.

The ohm has in the past been determined

in absolute measure using a specially con- structed coil system of known geometry. The

mutual inductance of such a system may be

determined from its geometry and dimensions

using electromagnetic theory. A commutated

dc method may then be used to obtain the

30
value of a resistance as a linear function

of the mutual inductance (length) and the rate of commutation (time). The resulting

overall uncertainty of the last experiment

of this type performed at NBS in 1948-19^9

was 10 parts per million T2] - rather large

especially when compared to the precision

with which resistance standards may be intercompared (ML01 ppm) and their drift rates relative to one another (^0.05 ppm/year).

More recently, it has been determined using

a calculable cross capacitor. This device

is based upon the theorem of Thompson and

Lampard which describes geometric conditions

for the construction of a device whose ca- pacitance is a known function of the displacement of a moveable electrode and hence

can be calculated if the displacement can be

measured. A guadrature bridge is then used

to effect a calibration of resistance in terms of capacitance. This experiment thus

establishes both the farad and the ohm in absolute measure. This is a much more ac- curate experiment than either the previouslydescribed ohm determination or the ampere determination, having an uncertainty of 0.06

ppm [3].

The volt may also be obtained in absolute

measure. Experiments are now underway at NBS and in the national laboratories of

several other countries. The NBS experiment

involves the measurement of the dc voltage

applied to concentric cylindrical capacitors

in terms of the electrostatic force generated between them as a function of their

relative displacement and the charge in ca- pacitance resulting from the displacement.

The target accuracies of these experiments

is of the order of 1-3 ppm. When completed,

these determinations will not only provide

the difference between the maintained and SI units of voltages, but when coupled with the

results of the Farad-Ohm determination, will

constitute an indirect determination of the

SI ampere for comparison with previous directly-obtained values.

The legal units are maintained at NBS in the form of assigned values to groups of

artifact standards. The legal farad is embodied in a group of ten, very stable,

fused-silica capacitors [4]. The mean ca- pacitance of the group has been assigned

from absolute capacitance determinations.

The legal ohm is the mean 4-terminal re- sistance of a group of Thomas-type standard

resistors immersed in oil at 25° C under a power dissipation of 0.01 watts. This value

is slightly inconsistent with the results of

absolute experiments and will be adjusted

after further work. It is not anticipated

that this adjustment will be as large as one

part per mil 1 ion.

The ampere, however, cannot be conveniently maintained using a physical standard.

Instead, the absolute measurements of resis- tance and current are used to derive an absolute volt from which a legal unit of voltage may be assigned. Until 1 97 2 , the volt was maintained 'via a group of saturated

standard cells in a temperature-controlled

enclosure. Since July 1, 1972, use has been

made of the ac Josephson effect to maintain

the legal unit. The ac Josephson effect is

a superconducting, pair-tunneling phenomenon

- the discovery of which has permitted the

construction of a linear f reouency-to- vol t- age converter whose proportionality constant

is invariant with respect to time and other

factors at the part in 10° level [5]. Figure 1 depicts the derivation of the basic

electrical units. It does not show an ex- periment for the direct absolute determination of the volt presently underway. Table

2 gives characteristics of the artifacts

embodying the leqal units. Appendix B con- tains a description of both the primary

electrical standards and the working stan- dards used to provide for the dissemination

of the electrical units.

The other quantities listed in Table 1 are derivable at various levels of accuracy

over their ranges. Accuracies of calibrations of standards in each of these areas

are more than adeguate to meet most static

measurement requirements except the followi ng

:

(a) Capacitance between 30 kHz and 1 MHz,

needed to support measurements resulting

from increasing carrier frequencies in com- munications systems.

(b) AC voltage from 0.1 Hz to 100 Hz at millivolt levels and below for the support

of vibration measurements aboard helicopters

(c) DC conductivity measurements in fer- rous and nonferrous metals for the support

of nondestructive testing of airframes and

other structures.

The standards and capabilities mentioned

to this point apply only to static measurements, i.e., those measurements for which

the time of measurement is not a constraint.

The advent of integrated circuits and the

resulting enormous increase in complexity of

electronics has resulted in reguirements for

many more measurements in industry to ensure

the quality of products and for maintenance

purposes. Manual measurement methods are totally inadequate to the demand. Fortunately this same new technology has provided

the system with the tools needed to perform

measurements at very high speeds - fast settling instrumentation to convert analog elec

trical parameters to encoded digital signals

high-speed data handling equipment for routinn digital and analoo signals, and readily

available inexpensive digital computers to control experiments, iollect data, direct it

31
CLOJ E <_ O

S- 01 3 U- O

CD •!- U JO

n3 fO

u a

<4

L_':__ k—j

r

i 4

i

i

v |

i 6

32
Table 2. Characteristics of the basic legal units.

Nominal Maintenance Absolute

Unit Artifact Value Accuracy Accuracy

Ohm Thomas-type resistor 1 ohm 0.08 ppm .06 ppm

Volt Josephson apparatus 10 mV 0.1 ppm 7.8 ppm

Farad Fused-silica capacitor 10 pF 0.4 ppm 0.034 ppm

Henry Maxwell -Wein bridge 200 ppm 200 ppm

Maintenance accuracy is that with which a perfect standard of equal

magnitude could be calibrated.

^Absolute accuracy is the accuracy with which the difference between

the SI and legal units has been reported.

to storage devices, and perform the mathematical manipulations necessary to put it in useful form.

Unfortunately, standards to provide a basis of accuracy for these high-speed or dynamic measurements simply do not exist.

There is no common, well-defined terminology

for describing the performance of these modern measurement tools, nor are there ac- cepted test methods or artifact standards

for verifying their accuracy under use conditions. There is no reliable way of precisely determining the trade-off between

speed of measurement and accuracy. Presently, dynamic measurements are supported

through the rapid measurement of static standards, a practice which does nothing to verify the capability to measure the rapidlyvarying parameters characteristic of modern

electronics. Other time-related properties

of the measurement instrumentation such as the rate at which they should sample, the

frequency of the measurements made, or delay

times after switching are determined on a trial-and-error basis. Such standards as uo exist are the tntracompany standards o* each

of the instrumentation and module companies

and, while they serve to achieve a degree of

uniformity within each company's product

line, are not directly relatible on an intercompany basis and certainly not available to the customers of the companies. In fact,

such standards are considered proprietary in that they seriously af^ec* product development in one of the most competitive marketplaces in existance.

This lack of standards and measurement

verification capability results in undue

costliness of measurement equipment, in hardware, testing and design costs, and in un- fulfilled measurement capabilities. Undue

cost accrues as the result of overspecif ication of components and modules at design

time. In addition, validation procedures

must be developed by trial-and-error techniques where adquuate standards do not exist.

A program for the development of standards

and test techniques to support the dynamic

measurement of electrical parameters has re- cently been initiated within the NBS. It is visualized that the hardware and written

standards developed in this program will be

made available to manufacturers and users of

electronic measurement equipment in the form

of calibration services, measurement assurance programs, and publications. As such

they will provide the basis needed for uniform and compatible dynamic measurements

throughout the country in the same way that

existing standards now do for manually-made

measurements

.

2.2 Basic Technical Infrastructure

2.2.1 Documentary Specification System

2.2.1.1 Standardization Institutions

There ?rp K *sically two major international standardizing organizations impacting

the Electrical Measurement System. The In- ternational Electrotechnical Commission (IEC)

is concerned with the production of standards, terminology, and specifications in support of electrotechnology. It has ap- proximately 75 technical committees to perform this function. A list of those technical committees most affecting the system is given in table 3. In this country, a delegate to each committee of interest is ap- pointed by the chairman of the national com- mittee. The delegate, in the case of each

proposed standard, attempts to develop a consensus U.S. position by getting technical

support from interested parties in industry

and government. The work of the IEC is very

important to U.S. foreign trade activities.

If standards which are essentially incompat33
ihle with common American practice are developed, they could be used aqainst the U.S.

in restraint of trade. More importantly,

standards developed overseas tend to be re- strictive in two other ways. In many cases, methodology is stressed rather than performance. This tends to inhibit technological

advancement in some product areas. Furthermore, the level of technology reflected in some standards is low by comparison to that

in the U.S. These two factors can readily

couple to close foreign markets to U.S. manufacturer',.

Table 4. 0 IML Pilot Secretariat 15

Tap 1

1

I r

T committees affecting the

Electrical Measurement Svstem

Committee Title

1 Terminology

3 Graphical Symbols

13 Measuring Instruments

25 Quantities and Units, and

their Letter Symbols

Instrument Transformers

42 High Voltage Testing Techniques

45 Unclear Instrumentation

50 Environmental Testing

56 Reliability and Maintainability

62 Electrical Equipment in Medical

Practice

66 Electronic Measuring Equipment

The International Organization for Legal

Metrology (CIML) is a recently-formed group

concerned with legal aspects of metrology,

such as weights and measures. Within 0 IML

is a pilot secretariat, Secretariat 13, for

the measurement of electric and magnetic

quantities which is concerned with the pro- vision of traceability for those kinds of

measure!: ents presently legally required.

Secretariat 13 is held by the U.S. and has

the subcommittees listed in table 4. Within the U.S., there are a wide variety

of legal and regulatory standards which impact the Electrical Measurement System. In the electronics world, traceability reauirements for suppliers are made mandatory

through contract by the federal government's

largest consumers: the Department of Defense,

the Department of Transportation, and the

National Aeronautics and Space Administration. The Department of Energy requires

that all contractors running installations

for them have a stylized quality control system, and they provide for the servicing of

those systems through their laboratory in Albuquerque, Nr\ run by the Sandia Corporation.

.Measurement of Electric and

Magnetic Quantities

Subcommi ttee

13.1

13.2

13.3

13.4

13.5

13.6

13.7

Title

International Compatibility

of Primary Etalons

Power Meters (wattmeters)

Energy Meters for Direct

Connect i on Energy Meters Designed to

be Used With Measurement

Transformers

Measurement Transformers

Indicating Instruments for

Voltage, Current and

Frequency

Recording Instruments for

Voltage, Current and

Frequency

The Department of Defense contractually

requires of its suppliers compliance with

military specifications MIL-Q-9859A, Quality

Program Requirements; MIL-I-45208A, Inspection System Requirements; and MIL-C-45FP2A,

Calibration System Requirements. By doing

so, it forces manufacturers to set up calibration and quality control systems which

require periodic calibration of all measurement equipment, traceability to national

standards in the case of all such calibrations, complete documentation of the calibration system including test procedures and

traceability, and any instrument undergoing

calibration to be tested using standards at least four times more accurate than it is. NASA has a similar but more rigid set of

regulations - NPC 200-1 A , Quality Assurance

Provisions for Space Contractors. The De- partment of Transportation requires traceability in its highway safety test programs

by specifying compliance with provisions of

the Society of Automotive Engineers' Standard J211b.

Of great potential impact are new and

forthcoming requirements of the Occupational

Safety and Health Administration (OSHA).

That organization has proposed that all industrial electronic and electrical equipment

comply with certain minimum safety standards.

Furthermore, it is proposed that all devices

be tested for compliance in certified testing laboratories and that tests must be con- ducted in a laboratory not belonging to the

manufacturer of the item being tested. At present these proposals are being appealed

by industrial groups.

34
The electric power industry subset of the

measurement system is also affected by the

mandatory safety requirements of the Occupational Safety and Health Administration.

Safety testing will be required of apparatus

used in work environments. Once again it is likely that all testing will be done by ac- credited laboratories.

At present, the Food and Drug Administration fosters safety standardization in the

manufacture of medical devices and instrumentation by sponsoring and promoting the

generation and adoption of standards with

the voluntary cooperation of industrial concerns, medical experts, and their own experts. Recently (May 1976) enacted and

signed into law is The Medical Device Safety

Amendment to the Food, Drug and Cosmetic

Act, which will set up control mechanisms

for medical instrumentation administered by

the Food and Drug Administration. These

mechanisms will range from having to obtain

approval before marketing to simply having

to register products in such a way as to enable an effective recall process in the

event defects are found. In addition, manufacturers will be subject to a number of

standards called Good Manufacturing Practices. These Pood Manufacturing Practices,

about to be promulgated, will require the

existance of formal quality control procedures (in many cases where none exist now)

and, in some areas of critical application,

the establishment of formal calibration

support systems with measurements required

to be accurate in terms of national standards. This will result in the creation of new standards laboratories and perhaps in recognition of new types of measurement

appl ications.

The bulk of the regulations affecting the

electric power industry come from state utility regulatory commissions. These com- missions have broad powers in regulating the

operations of electric utility companies.

They approve rates and siting of power plants

and major transmission lines; to some extent

they have a jurisdiction over the quality of

electric power service—occasionally they

prescribe tolerances for the supply voltage';

they regulate the accuracy of watthour meters and the quality control systems which

have been set up to test both newly-acauired

and previously-installed meters. There are a large number of voluntary

standards organizations impacting the Electrical Measurement System. The majority are

listed in table 5. Those organizations

whose titles are starred have the greatest

impact since they produce standards which

have affect on the higher accuracy measurement processes, instruments, and artifact

standards which support the system. MBS

personnel participate in the activities of a number of committees sponsored by each of

these organizations.

Table 5. Voluntary standards sponsoring

organi zations

Aerospace Industries Association

American Home Lighting Institute

American National Standards Institute

American Society for Quality Control

American Society for Testing and Materials

American Welding Society

Association of American Battery

Manufacturers

Association of Edison Illuminating

Companies

Audio Engineering Society

Edison Electric Institute

Electric Apparatus Service Association

Electronic Industries Association

Institute of Electrical and Electronic

Engineers

Institute of High Fidelity

Institute of Printed Circuits

Instrument Society of America

National Association of Relay Manufacturers

National Electrical Manufacturers

Association

Radio Technical Commission for Aeronautics

Resistance I'elder Manufacturers Association

Scientific Apparatus Manufacturers

Association

Society of Automotive Engineers

Underwriters Laboratories

Variable Resistive Components Institute

5f - Organizations having major impact

2.2.1.2 Survey of Documentary Standards

A survey on documentary standards affecting the electrical measurement system was performed. Consideration of documentary

standards in the electrical area shows that

the documents may be divided into six cate- gories: safety specifications and testing

methods, terminology, dimensional requirements for compatibility, testing methods to determine electrical properties, performance

parameter specifications and testing, and

reliability assurance procedures. It is ap- parent that all except dimensional requirements involve or affect electrical measurements in some fashion. Since there was no evident valid means of ranking or nuantifying these various effects, and since any

document directly concerning etalons (artifact standards), measurement apparatus, and

instrumentation would be more likely to have

a large impact on the measurement system,

those documents were considered separately.

35
Investigation of documents sanctioned by

the International Fl ectrotechni ca 1 Commission

showed that, of the population of 759 standards and supplements, 45 referred directly

to etalons, instrumentation, and precision

measuring equipment. Most of these specify

moderate accuracy instrumentation. Of the

remainder, perhaps ten percent deal with

terminology and symbols and another ten percent with dimensional specifications. As has been pointed out, the remainder have a bearing on the segment of the system dealing

with the testing of export products for com- pliance with these standards to assure ac- ceptance abroad. And, although a third or more of the remaining documents refer to radio frequency measurements above 1 MHz and

therefore outside the scope of this study,

measurements at those frequencies reguire a degree of low frequency measurement support

[6].

A similar investigation of standards issued by the American National Standards In- stitute and having bearing on electricity and

electronics, revealed that, of ndd standards,

supplements, and handbooks issued, 20 provided definitions, specifications of performance, and test and verification methodology

for standards, instrumentation, and precision measurement apparatus. Terminology is largely concentrated in 0^2.100, the Institute of Flectrical and Electronic Engineers

(IEEE) Standard Dictionary of Electrical

Terms. About ten percent of the documents

in this study specify dimensions for compatibility. The remainder specify or outline

tests or measurements of various electrical

parameters. The standards sanctioned by

this institute are perhaps the most universally accepted in this country. A very high

percentage of them have been adopted directly from the work of other groups such as the

IEEE, the National Electrical Manufacturers

Association, and the American Society for

Testing and Materials [6]. mary standards

Generally speaking, IEEE standards dealing directly with measurement apparatus,

instrumentation, and etalons have been

adopted by the American National Standards

Institute and, hence, are covered above.

Standards and recommended practices produced

by the Instrument Society of America deal

primarily with the application of instrumentation to industrial processes and with the use and measurement of transducers. There

are 18 such procedures involving electronic

instrumentation. Standards produced by the

Electronic Industries Association deal primarily with electrical testing of components

and with electronics in the Manufacturing

process

.

The Edison Electric Institute, the As- sociation of Edison Illuminating Companies,

and the National Electrical Manufacturers

Association have contributed to standards

for electric power systems, either through

sponsorship of, or participation in, ANSI

committees. In the measurements field three

standards are very prominent: ANSI C12

(watthour meters, electricity metering; com- mittee is chaired and co-sponsored by NBS);

ANSI C57.13 (instrument transformers); ANSI

C93.2 (coupling capacitor voltage transformers) [7]. The IEEE is a very important

contributor to the measurement and test standards for electric power systems through the Power System Instrumentation and Measurements (PSIM) Committee and its subcommittees

and working groups. This committee is re- sponsible for about 10 standards, most of

which have been adopted by ANSI.

There are two areas in which the standards reviewed have little or no impact.

The first is the area of the dynamic mea- surement of electrical processes in which

timing is a significant factor. With the

increasing use in this country of computerdriven measurement equipment, standardization of terminology and specification of

valid timing test methods are necessary to relieve the chaos which presently exists in the marketplace for that type of equipment

[6]. (See also Appendix C.)

Methodology for testing the measurement

system is not well documented as such. Although documentation of tests on certain

types of instruments is available, technigues for the verification of the measurements are not documented. No mention is made in published standards of the type of

system-output sampling that is employed in many of the Measurement Assurance Programs

being carried out by the NBS. The importance of these voluntary standards is reflected by the support given to their generation by industry and the government. Most standards committees meet at least biannually and the meetings are generally well attended even though the participants are from various parts of the country.

And, although the use of these standards in this country is on a voluntary basis, their

generation nives a solid technical base for

U.S. input for international standards work,

the results of which may carry the force of

law in other countries. Heavy and successful U.S. activity in that area is needed to enhance our export trade posture.

2.2.2 Instrumentation System

2.2.2.1 Measurement Tools and Techniques

Practical (end use) measurements in the

Electrical Measurement System are supported

by a hierarchical structure resembling an inverted pyramid with the National Bureau of

36
Standards as its base. At NBS, the legal

electrical units, which have been determined

in terms of the theoretical (SI) units, are maintained and disseminated to the standards

and calibration laboratories ot the community. (See Appendix E for a list of NBS primary and working electrical standards.) The

units are transferred there from standards

to a class of instrumentation commonly known

as test equipment. It is this test equipment which provides the means for the calibration and maintenance of virtually all electronic and electrical equipment in the

country, regardless of its nature.

A few basic types of standards, used in corporate standards and calibration laboratories and supported by NBS, are used to tie

the units embodied in test equipment to the

legal units. Table 6 lists these basic standards. Standards are arranged in descending

order of performance capabilities where more

than one are listed. Approximate accuracy

capabilities are expressed in parts per million (0.0001/c) in parentheses.

In parallel with the hierarchical organization mentioned above, there are similar

technical chains of measurement traceability

from NBS standards to the end uses. ExamTable 6. Basic units and industrial standards

pies of these are to be seen in figures 2 and 3 which depict the traceability of ap- plied voltage measurements in the electronics industry and applied energy measurements

in the electric power area, respectively.

The dissemination of electrical units is generally accomplished in one of two fashions. In the traditional calibration method, the standard to be calibrated is taken

to the laboratory in which the standard used

to calibrate it is kept. It is then calibrated, and returned. Although this ap- proach has the merit of being easily managed

and is generally adequate in most situations,

it has a number of technical flaws. The

standard is not calibrated in the environment in which it will be used. Hence possible environmental effects tend to be overlooked. The same holds for the electrical

conditions of use. Checks are not made on the veracity of the measurement made in the use of the standard; and no assessment of

the stability of the standard during the

physical transfer is generally made. Alternatively, the units may be disseminated by

using suitably tested transport devices to sample the quality of measurements being

made in the measurement chain. This latter

Unit of

DC voltage

DC resistance

DC resistance

ratio

Capacitance

Starting

Level

1.02 V

10

n * ohm

1/1

10 pF

Standards

Inductance 100 mH

AC voltage ratio N/A

AC-DC voltage N/A

difference

Power N/A

Energy N/A

AC current ratio N/A

Saturated standard cell enclosures (0.5) ; solid state reference devices (5), unsaturated standard cells (50)

Primary standard resistors (0.1-1); Rosa-type and other

intermediate standard resistors (1-15); resistance

transfer boxes and precision decade resistors (10-250);

resistive shunts (50) Hamon devices (0.02); precision resistive dividers (0.05);

standard volt boxes (5); potentiometers (10)

Fused-silica dielectric capacitors (0.4); gas dielectric

capacitors - 3-terminal (3-100); g?s dielectric capacitors

- 2-terminal (50-1 ,000)

Air core standard inductors (200-2,000)

Inductive voltage dividers (0.1); instrument transformers

(100-300); high voltage transformers (100-3,000)

Multifunction thermal voltage converters (5-10);

Hermach-type thermal voltage converters (5-100)

Wattmeters (100-10,000)

Watthour meters (200-1,000)

Current transformers (100-300)

t Numbers in parentheses are approximate uncertainties in ppm.

n can be any whole number between -4 and 14.

37
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38
Electricity Division 1

J Time & Frequency

Division

Time

1. Standard Cells

2. Potentiometer

3. Standard

Resistors

4. Volt Boxes

5. Voltmeter

Primary Energy Std.

(3 watt-hour meters)

Current Transformer

Voltage Transformer

Transport

Watt-hour

Meter

Field Standard

(Rotating watthour meter)

New watthour meter

(sampl ing or

100% testing)

Installed

meter

(sampl ing)

Figure 3. Traceability of energy measurements

39
method is somewhat more complicated in terms of data analysis and management and more

costly in labor, but gives a more realistic

picture of system performance. This approach

is now being used more and more for dissemination purposes, especially where high ac- curacy is desired for the dissemination of

the basic units between primary laboratories,

or in cases where a suspicion exists that a measurement process is in trouble.

Technical information is disseminated

throughout the system in the following ways:

(a) written calibration procedures,

(b) manufacturers' instruction manuals

and user newsletters,

(c) informal contacts between colleagues,

(d) technical papers published in the

open literature, and

(e) voluntary standards (previously discussed).

Most information is probably disseminated

through the use of published procedures.

Most government quality control contractual

requirements demand the establishment of a library of written procedures covering everything from data handling to instrument calibration. Such procedures are exchanged

throughout the segment of industry supporting the government through GIDEP, the Government Industry Data Exchange Program. Participation in this data-bank program is now mandatory for Defense Department contractual

suppliers. Procedures must be submitted at

a given rate by each participant for inclusion in a giant data bank. The bank is ac- cessible to all participants. (GIDEP is not confined to calibration information. Other

aspects include reliability data, instrumentation applications, etc.)

Many manufacturers include calibration

instructions as well as application instructions in the user manuals supplied with instrumentation. This is probably the second

major source of technical information in the

system. The others seem to be less significant.

2.2.2.2

The instrumentation industry [8,9] is the

only source of those measurement tools which

provide the foundation for the quality of

all manufactured products and, in some cases, provide the enabling means of their manufacture. Instruments are manufactured to measure every electrical parameter with accuracies ranging from those to be found in

standards laboratories (parts per million)

to five percent for analog panel meters. Their use with transducers enables the mea- surement of other physical parameters, such

as temperature, strain, force, pressure,

resistance, etc., as well. Instruments can

be as simple as the dwel 1 -tachometer used to tune an automobile or as complicated as a system for the checkout of inertial navigators, employing dozens of separate pieces of

test equipment, capable of 0.005% accuracy

in some cases and completely under the control of a computer.

Measurement instrumentation and control

equipment enable the large volume, high

quality, and low cost mass production which

has made our high standard of living

possible. The ready availability of digital

computers as well as of reliable, accurate

instrumentation capable of being computerdriven has led to the increasing use of electronic process control systems in areas such

as the petrochemical industry to replace ex- isting pneumatic, hydraulic, or manual systems. Electronic systems have the virtue of

being able to respondmuch more rapidly - nearly instantaneously - to process changes

than mechanical systems and offer greater

flexibility in that process modifications

may be readily brought about through soft- ware rather than hardware changes. In addition, the use of digital logic makes

availabile a capability for decision-making

of a complex nature. Electrical measurements are necessary to the maintenance of this type of electronic

equipment as well as all others of a durable

nature.

A representative list of electronic instruments used for calibration, maintenance,

testing, and analysis is given in table 7. Table 8 is a list of the major manufacturers

of basic standards and primary calibration

equipment in common use in standards laboratories in the U.S. It is not exhaustive.

The Electronic and Electrical

Instrumentation Industry

40
Table 7. List of common instruments

Acoustic noise generators

Admittance meters

Ammeters

Amplifiers (over 30 types)

Analog-to-digital converters

Audio frequency analyzers

Audio mixers

Audio test oscillators

Automotive test systems

Bridges (17 types)

Calibrators (current, voltage,

oscilloscope)

Current sources

Data recorders

Decade boxes (capacitance, resistance)

Digital comparators

Digital filters

Digital-to-analog converters

Digital voltmeters

Distortion analyzers

Electronic integrators

Electronic micrometers

Electrostatic voltmeters

Fl uxmeters

Function generators

Harmonic analyzers

Hysteresis plotters

Interferometers

Instrument transformers

Multi tester meters

Operational power supplies

Oscil loscopes

Peak power meters

Potentiometers

Refl ectometers

Sampl ing vol tmeters

Tachometers

Telemetry modulators

Vibration meters

Voltage-to-frequency converters

Volt ratio boxes

Watthour meters

Wattmeters

Table 8. List of electrical standards

manufacturers

Beckman Instruments, Inc. James P. Biddle Co,

Electro Scientific Industries

Engelhard Industries

Eppley Laboratories

Peneral Electric Company

RenRad

Guildline Instruments Ltd.

Hewlett-Packard Company

John Fluke Manufacturing Company

Julie Research Laboratories

Keithley Instruments

Knopp Inc. Leeds and Northrup Company

Penn Airborne Products

Sangamo Electric Company

Ti ttex Victoreen

Western Electric Company, Inc. Yokagawa Electric Works

Test equipment and instrumentation manu- facturers constitute a much larger segment

of industry (SIC Codes 3811, 3821, 3822, and

3825). Total estimated sales in these cate- gories for 1976, according to the Domestic

and International Business Administration,

U.S. Department of Commerce, were in excess of $7.7 billion [10]. Domestic customers

included every segment of industry using

high technology.

Electrical measuring instruments shipments were estimated to value $2.1 billion

in 1975 and are expected to increase to $2.4

billion in 1976 [10]. Test instrumentation

accounts for 60 percent of the output of

this SIC (3825). Low accuracy meters, used

in panels and switchboards, and portable

meters constitute 28 percent, and watthour

and other integrating meters make up the re- mainder. The test equipment industry is a high technology industry which is noted for

rapid exploitation of technological advances.

Some manufacturers spend as much as ten per- cent of their annual sales on research and

development [91.

The growing semiconductor market, in- creases in the complexity of components, and

their growing application to consumer products has generated an expanding market for

automatic test equipment for production line

quality testing and adjustment. Automatic

test equipment is increasincily being used by

semiconductor device users as well as manufacturers. Digital instruments are rapidly

replacing analog equipment due to substantial cost reductions and the minimization of

reading error.

41
The instrumentation market is expected to expand over the next few years as manufacturers are spurred by the need to conserve

energy and costs and reduce pollution.

In the category of Engineering and Scientific Instruments (3811), navigational and

guidance instruments, and automatic pilots

for ships, aircraft, and space vehicles com- prised nearly one half of the industrial output. Thirty percent was composed of laboratory instruments and equipment. The remainder was composed of engineering apparatus for

meteorological, hydrological , geological,

civil, and mechanical engineering purposes.

Production of measurement and control instrumentation was expected to reach $2.8 billion

in 1976 [10]. The process control portion

of this industry is the portion to be con- sidered part of the electrical instrumentation industry. There is a high degree of

activity in this area spurred by manufacturers' demands for processes which are more

economical or efficient and which have re- duced energy consumption and environmental

impact.

Exports of all instrumentation are presently at the $60.1 million level [10], having risen sharply over the last two years.

Although the U.S. holds a strong lead in the most advanced instrumention, foreign competition is increasing. The recent increase

in exports is expected to moderate somewhat

in the future and imports are expected to increase si ightly [9]

.

2.2.3 Reference Data

Some reference data, i.e., physical con- stants and other similar data, are generated

in the NBS and the academic portion of the

measurement system and used primarily by re- search scientists seeking to test and improve

physical theory. Examples of such data are measured values for -yp, the gyromagnetic

ratio of the proton, and e/h, the ratio of

electronic charge to Planck's constant [5,

11], both obtained experimentally at NBS. Recent accurate measurements of the speed of

light have made more accurate determinations

of the absolute ohm possible [3,11].

Although their use throughout the bulk of

the measurement system is limited, their determination has a fundamental impact on the

values of the basic maintained electrical

units and as a source of improved methodology

for monitoring the stability of those units.

Furthermore, much work has been done at NBS to supoly least-squares adjusted values

for the fundamental physical constants [12].

These values, internationally sanctioned by

CODATA, the Committee on Data for Science

and Technology of the International Council

of Scientific Unions, tie together all of

the remote branches of physics and are universally used by theoretical and research

physicists. Data of this type are disseminated by publication in the scientific literature.

A more widely used type of reference data

is that of electrical performance characteristics used by all design engineers. This

type of data is disseminated by publication

in various engineering and scientific handbooks, prepared by technical institutions

and private corporations, such as those given

in table 9. Data in these handbooks range from con- ductivity of wire, breakdown characteristics

("volt-time curves") for sphere gaps, and

Kerr electro-optical coefficients to recom- mended circuits for various applications.

The most universal of them are obtained as the result of experiments done in national

laboratories such as NBS, and in universities. Others, such as transistor perfor- mance characteristics, are published by semiconductor device manufacturers. All are es- sential to the efficient functioning of the

design engineer and the experimentalist. It

is apparent that if these data were not generally available, each new advance in technology and each design of a new circuit would

come only after a long period of trial and error, measurement and experimentation.

2.2.4 Reference Materials

The direct role of reference materials in the Electrical Measurement System is limited

to two areas: those of eddy-current measurements of the conductivity of non-ferrous

metals and measurements of resistivity of

silicon used in the semiconductor industry.

The eddy-current technique is widely used in the aircraft industry and its metals sup- pliers as a non-destructive means of determining the heat treat of alloys, notably

those of aluminum and titanium. Such mea- surements are commonly made using eddy-current conductivity meters which actually measure the change of inpedence of a probe,

driven at a certain frequency, caused by its

ccuplinq with a conducting material. Such

meters are calibrated using small standard

samples (coupons) of metals manufactured

either by Boeing Aircraft Corporation or the

manufacturers of the meters. The conductivity of the coupons is generally determined

in terms of the legal unit of resistance

using dc measurement methods performed on rather large samples of materials which are

then cut up. However, dc measurements thus made are of

the average conductivity through the entire

metal piece and in one direction only. Eddycurrent measurements are localized in a small

42
Table 9. Sample sources of reference data for engineers

Source Publ i sher

Reference Data for Radio Engineers

Handbook of Chemistry and Physics

The Semiconductor Data Book

Formulas for Computing Capacitance and

Inductance, NBS Circular 594

American Institute of Physics Handbook

Digital Logic Handbook

The Transistor and Diode Data Book

International Telephone and Telegraph

Corporation, New York, MY Chemical Rubber Publishing Co.,

Cleveland, OH Motorola Semiconductor Products,

Phoenix, AZ

National Bureau of Standards,

Washington, DC American Institute of Physics, New York, NY Digital Equipment Corp., Maynard, MA Texas Instruments, Dallas, TX

volume in contact with the probe and usually

limited to a shallow depth. Because of the

structure of most metal samples, (grain orientation, impurities, local strain, etc.)

the local conductivity can vary widely and

intercomparisons of eddy-current meters and

conductivity coupons have revealed some ra- ther large discrepancies in measurements

being made. Accordingly, MBS has been approached by a number of aircraft companies

to establish material standards in order to eliminate these discrepancies and to remove

the confusion which exists in this area. In 1977, NBS held a workshop in this area to identify the problems associated with these

measurements and identify other areas where

standards are needed. Plans have been made

and work is underway to establish national

standards of conductivity in the range be- tween 1% and 100% of the conductivity of

copper. Based upon this work, eddy-current

conductivity standards or coupons of various metals will be made available through

the Standard Reference Materials Program.

Standard reference samples of silicon are available through the Standard Reference

Materials Program to support the measurement

of the resistivity of semiconductor materials. This SRM results from work done by

the NBS Electronic Technology Division and

will be reported in that Division's impact

report.

2.2.5 Science and People

Electrical metrology involves the direct

application of principles of physics and

electrical engineering to the solution of

measurement problems. The role of physics

is most evident in basic metrology work; the

development of instrumentation and the as- sembly of sophisticated instruments into measurement systems. The major contribution of

the measurement system to its undergirding

science and technology is to provide a common means of communication, a universal

technical language without which the transfer of knowledge and technology within the

system would be impossible. Without such

data transfer, the advanced technological

society we find in the United States would

not be possible.

There is heavy involvement of professional societies in the Electrical Measurement System. They provide a means of informal and formal communication of knowledge

throughout the system through their meetings

and publications. The following is a list

of these societies:

1. Institute of Electrical and Electronic

Engineers; Group on Instrumentation

and Measurement

2. Instrument Society of America; Metrology

Division

3. Precision Measurements Association

4. National Conference of Standards

Laboratories

5. American Society for Quality Control

6. American Society for Testing and

Material

s

7. American Physical Society

8. IEEE Groups on

a. Automatic Control

b. Biomedical Engineering

c. Circuits & Systems

d. Communications

e. Industrial Electronics and Control

Instrumentation

f. Reliability and Maintainability

9. IEEE Power Engineering Society

The first four of these are groups dealing directly with electrical metrology and

its applications. Their members produce

document standards, engage in measurement

activities, such as round-robins, and, most

importantly, communicate and encourage com- munications by means of meetings, colloquia,

conferences, seminars, and producing various

43
publications. The regaining are trore peripherally involved in that their activities

are supported by the system and they provide

the system with backup technology and information.

The major electrical metrology publications commonly used in the U.S. are:

Metroloaia

IEEE Transactions on Instrumentation

and Measurement

IEEE Transactions on Power1 Apparatus and

Systems

Measurements and Data

Ins trumentati on Techno 1 ogy

Review of Vaienttfia Instruments

All feature technical papers on metrology

in various application areas or sectors of

the measurement system. Pore specific, detailed information on specific calibration

problems is disseminated through documented

calibration procedures as discussed previously.

Measurements people generally fall into one of five categories: physicist, electronic engineer, electrical engineer, statistician, or electronic technician. Those in the

first four receive their professional training in universities and colleges. Electronics technicians generally receive their education as the result of military service and

occasionally through correspondence or technical schools. All can improve or augment

their background through the many seminars

and short courses in metrology and electronics offered around the country. The

field of metrology is fortunate in having a very high degree of dedication on the part

of its workers.

2.3 Realized Measurement Capability

The graphs on the following two pages

represent measurement accuracies available

throughout the measurement system. It must

be understood that they represent an over- simplification of an extremely complex interacting set of parameters. For example, the

question of a specified time frame in which

a measurement is made has been ignored. En- vironmental and power line effects have been

likewise ignored. The fact that a good metrologist can coax performance far in ex- cess of manufacturers 1 specifications from

measurement instrumentation, given appropriate resources or complete freedom of

choice regarding methodology, is not represented. In essence, the graphs represent a compilation of manufacturers' specifications

and known behavior traits of populations of

standards in common usage. In the interest

of brevity, ac quantities have been shown

at one freauency, 1 ODD Hz in the case of capacitance and inductance, fO Hz in the case of

energy and current ratio, and 10 kHz in the case of rms ac voltage.

The instrumentation available is adequate

for the fulfillment of static measurement

needs at the present time in most cases. Re- cent evidence, however, indicates the possible existance of a hitherto unsuspected

loss of measurement accuracy as a result of

high accuracy electronic instruments having

been physically transported, either to be

calibrated or for use elsewhere. The problem is presently being investigated.

Another problem area, whose importance is emphasized in liaht of present economic con- ditions, is the cost of maintaining an instrument in calibration. Many new instruments

are being designed with self-checking features

to permit extension of calibration intervals

or the elimination of periodic calibrations.

A method of testing the verity of self-test

features is desperately needed.

More and more measurements are being made

by automatic test systems; assemblies of offthe-shelf commercial instruments, customized

apparatus, and interconnecting circuitry,

all controlled by computer. Generally such

systems have some self-test capability which

provides a functional check and perhaps a limited calibration using an internal standard as a reference. The self-test capability must be periodically augmented by

calibrations. Usually this is accomplished

by removing critical pieces of apparatus to

a remote laboratory for individual calibrations which are assumed valid when the

system is reassembled. Rarely are systems

calibrated as systems, despite the fact that,

because of temperature, interface, and electrical grounding and shielding problems,

they tend to be somewhat more than the sum of their parts. The major restraining fac- tors of systems calibration seem to be lack

of acceptance of the approach by military

and other government inspectors and the

nonavailability of hardware specifically

designed for the purpose. There does at present seem to be a growing awareness of

the advantages of this methodology.

There are a few areas in which present

measurement technology has fallen behind re- quirements. There is a problem, previously

discussed, regarding the use of samples of

nonferrous metals as conductivity standards

for nondestructive testing purposes. An- other area is that of ac voltage measurements throughout the audio frequency range.

Improved measurements in this area are needed to support modern ac voltage measurement

equipment. A third area is that of the mea- surement of impedance (capacitance) in the

frequency range between 10 kHZ and 100 kHz.

44
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The communications industry is in a trend

toward increasing carrier frequencies into

this range and have requested calibration

data there.

The energy crisis has created some high

voltage measurement problem areas. Because

of the limited supplies of petroleum, electric energy will comprise an increasingly

larger share of all energy consumed. Efforts are underway, sponsored by the Department of

Energy and the Electric Power Research Institute, to develop more efficient and safer

ways to transmit electric power. These efforts require new or improved methods and

standards for measurements of impedance,

ratio, voltage, and current at voltages as high as 1.2 million volts. They are needed

for the evaluation of dielectric losses in

insulators at cryogenic temperatures under

high voltage conditions, for measurement of

electric fields caused by ultra-high voltage

transmission lines (to test biological effects), and for the measurement of high voltage transient signals to test their effects

on lines, measurement apparatus, switchgear,

etc.

As noted, the entire section thus far has

concerned only static measurements, measurements in which time is not a critical factor.

However, the timing of a measurement is increasingly important for two reasons. Be- cause of the ever-increasing complexity of

electronic modules, components, instruments,

and systems, more and more testing is re- quired for quality control purnoses. As testing is the factor which limits the production rate of electronic goods, increasing

emphasis is being placed on "thruput".

Secondly modern electronic circuitry is more usefully characterized in terms of

transient responses to changing input con- ditions. This leads naturally to the need

to measure rapidly varying parameters, to be concerned with wave shapes as well as integrated energy content. This can only be

done by "sampling" or measurements at high

speed.

In order to increase the speed at which

measurements are made, a whole new world of

automatic testing has evolved. Analog-todigital converters with microsecond measurement times are computer-controlled and

switched in systems which permit thorough

testing of complicated circuits in relatively

short time.

Along with this evoluation has come a degree of chaos. The conclusions of a re- cently-held workshop on "Critical Electrical

Measurement Needs and Standards for Modern

Electronic Instrumentation" l"13] are:

(a) There is a need to introduce time as

a measurement parameter as a result of re- quirements of automatic test and control

systems.

(h) A host of new parameters have re- sulted from the application of automated

systems to the production line, parameters

for the measurement of which there are no standards, techniques, or even a common

terminology.

(c) A need for a new or higher accuracy

measurement has resulted from recent work

at the leading edge of measurement technology in electronics.

From these, a number of critical, specific areas needing immediate attention were

indicated. Among them were:

(1) capacitor and resistor characterization under pulsed conditions;

(2) dynamic performance characterization

for modern sianal conditioning and data con- version devices, such as aperature times of

sample-hold amplifiers, resolution-time

product for analog comparators, iitter error of digital-to-analog converters, etc.;

(3) characterization of settling times

for precision sources and measurement devices, for example digital voltmeters and

programmable power supplies;

(4) transport standards for system performance validation to be used as a basis

for Measurement Assurance Programs;

(5) effects of dynamic loading on sources

;

(6) prediction of long-term performance

of components and devices from short-term

behavior, for example, relating output noise

and device reliability;

(7) improved transportable dc standards

based upon solid state technology which would

be immune to short-circuit or loading damage,

not materially effected by temperature fluctuation, and perhaps at more useful levels

of voltage, current, or resistance than are present standards;

(8) measurement of dielectric hysterisis

in capacitors and other components; and

(9) generation of nonsinusoidal , high

crest factor precision signals for true rms

cal ibrations.

The workshop was attended by 2^ participants who represented a variety of industrial

and government organizations. The abstract,

preface, and list of participants from the

final report of the workshop (published as MBS Technical Note 865) are appended [Appendix C].

47
2.4 Dissemination and Enforcement Network

2.4.1 Central Standards Authorities

The Bureau International des Poids et Mesures (BIPM) is the supranational authority

of the electrical measurements world. In its

laboratories are housed artifacts embodying

international standards of voltage and resistance. At three-year intervals, BIPM undertakes the i ntercompa ri son of transportable

standards of voltage and resistance from the

major national laboratories of the world and

its own international standards. The published results of these intercomparisons are Generally accepted as quantifying the differences among the units of the participating

countries. BIPM also sponsors international

exchanges, such as a present round-robin intercompa ri son of transportable capacitance

standards. Such devices are measured at various national laboratories and the results

submitted to BIPM. At the conclusion of the

experiment the results are published.

The Consultative Committee on Electricity,

to which the Chief of the Electricity Division, NBS , is a delegate, advises and re- commends courses of action to BIPM in matters dealing with electrocal metrology.

The National Bureau of Standards is the

delegated legal authority for electrical mea- surements in the United States. Public Law

fil7, 81st Congress (21 July 1950), states in Section 12:

"It shall he the duty of the Secretary

of Commerce to establish the values of

the primary electric and photometric

units in absolute measure, and the

legal values for those units shall be

those represented by, or derived from

national reference standards maintained

by the Department of Commerce."

Most nations with modern technological ca- pabilities have organizations performing the

function of maintaining legal electrical

units. Aside from minimizing the effects of

discrepancies between legal electrical units

on export trade, the major interaction be- tween these laboratories and the Electrical

Measurement System is to provide an international redundance in the realization of the

absolute electrical units. Parallel experiments, perhaps with differing approaches and

performed under different roofs, offer the

best opportunity to eliminate the effects of

unidentifiable systematic errors from experimental results. Since adjustments to units

are made as the result of these experiments,

redundancy in them is of great importance.

2.d.2 State and Local Offices of Weights

and Measures

The Electrical Measurement System only

interacts with the weights and measures

field to the extent of providing the basis

for design and maintenance of any electronic

equipment used in that field.

2.4.3 Standards and Testing Laboratories

and Services

Three kinds of laboratories provide mea- surement services to the Electrical Measurement System. Standards laboratories are

those laboratories which can provide for the

calibration of electrical standards with

accuracies within an order of magnitude of

those offered by NBS. They may well be, and

usually are, capable of providing calibration services for nearly any type of instrument over any range of parameters and at any accuracv level up to those of the basic

standards. A calibration laboratory, on the other hand, calibrates and repairs test equipment and production-line equipment of more moderate accuracy. The third type of

laboratory is a testing laboratory. Although it may offer services similar to the

other two, it is characterized by facilities

for environmental, safety, and performance

testing.

The first two types can be either the

basic measurement laboratory of a large

corporation, the service repair department

of an instrument manufacturer, or a small

private company dealing only in those services mentioned. Standards laboratories

tend to be of the first two varieties,

whereas calibration laboratories tend to be

of the third.

Within the electric power industry one

can make a distinction between the utility

and manufacturing sectors. The major utility companies each operate a kind of laboratory which can be characterized as a mixture of the standards and calibrating

laboratory with some functions of a testing

laboratory. With respect to accuracy, the

most demanding reguirement is for watthour

meter and instrument transformer calibration. For these functions the laboratory

maintains high quality standards, with ac- curacies within an order of magnitude of

those of NBS standards. The same laboratory usually also calibrates and repairs

test instruments for the company. Finally,

the laboratory may have iurisdiction over

the acceptance tests of all incoming and

in-service watthour meters and instrument

transformers. The latter is a function of

a testing laboratory.

48
Within the manufacturing sector, the typical laboratory can be classified as a testing facility employed in quality control and

performance testing of a company's products.

This laboratory may obtain standards services

from a standards laboratory elsewhere within

the company, may occasionally ask for NBS

help, or may rely on traceability through the

supplier of the measuring instruments.

Appendix D contains a list, drawn from the sources indicated, of standards and calibration laboratories whose services are available to extra-company organizations. It is not exhaustive. The 13P laboratories listed,

along with others of their kind and the major

military and corporate laboratories supplying

services only within their own and subcontracting organizations, provide the traceability skeleton supporting the body of electrical measurements in this country.

Of federal laboratories other than NBS,

apparently only three offer electrical calibration services on a fee basis. They are:

(a) U.S. Army Electronics Command,

Ft. Monmouth, NJ

(b) Naval Air Rework Facility,

Norfolk, VA

(c) Navy Eastern Standards Laboratory,

Washington, DC

Other military and NASA laboratories will

generally provide a contractor with services

if they are not elsewhere available, or if

it is to the advantage of the government.

All the above-mentioned and those included

in the referenced appendix are laboratories

having services available to outside organizations on a fee basis. No mention has been

made to this point of the remaining group,

namely, those laboratories which provide services only internally, to their own organization. Of the 222 standards laboratories

listed in A Directory of Standards Laboratories [14], 164 have dc and low frequency

capability, and only 65 choose to make that

service available to outside firms or organizations.

An analysis was made in 1973 of NBS calibration clientele for the purpose of determining, among other things, the general magnitude of the Electrical Measurement System.

A representative sample of clients' names was classified by primary SIC code by Dun and

Bradstreet. The resulting list of code numbers was filtered to remove obvious measurement-void categories. Dun and Bradstreet's

data base, which contains information on over

3 million U.S. companies, was then searched

to provide information on all companies in

the relevant group with numbers of employees

in excess of 25. Information was provided

on 12,865 companies. Even though this figure

is not an accurate count of measurement-i nt ensive organizations due to the wav the individual SIC code numbers are assigned, it

is a good protrayal of the extensi veness of

the measurement system. In fact, the fiaure

is on the low side because companies engaged

in routine measurements without rigid trace- ability requirements are not included. Efforts to sample this grey area of the sys- tem have been minimal due to financial con- straints.

For companies not having rigid traceability requirements, the measurement system may

flow through the company via the electronic

maintenance shop or the outside organization

which performs this function for them. Many

of the laboratories in Appendix C also perform repair functions.

For organizations which must maintain

traceability because of contractual or legal

requirements, the flow of electrical units

through the system becomes more stylized.

The flow of the electrical units through

such an organization is depicted in figure

5. The corporate or primary laboratory may

receive its units from MBS or from any out- side laboratory. It is repsonsible for the

most accurate standards in the organization.

The actual organizational relationship of

these various components varies from corporation to corporation. An extremely large

corporation, such as Boeing Aircraft, may

have a corporate standards laboratory which

serves as a single tie between NBS and all

of its calibration laboratories. Other

large companies such as Honeywell and IBM have a standards laboratory at each location

and many such laboratories have standards

calibrated by NBS. Other corporations such

as 3M and Peneral Electric do not make the

use of any narticular structure mandatory.

Each opera ting division is autonomous and

may pursue its quality assurance program

quite independently.

In smaller organizations the standards

and calibration facilities are usually com- bined. Occasionally even the maintenance

shop is in the same organization, if not in the same room. These basic elements are

always present, however, even if not struc- turally distinct. Take the case, for ex- ample, of a small company employing about

poo persons, which is concerned with the

manufacture of temperature and air speed

transducers and allied instrumentation.

They have need of extremely accurate re- sistance standards because of their temperature work. However, their dc voltage re- quirements are not as strinaent. They have

their resistance standards calibrated at

NBS, but their corporate voltage standard is

a digital voltmeter calibrator which is calibrated for them periodically by another com- pany in the area. 49
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Representative organizational charts for

two larqe corporations and the U.S. Air

Force are to be found in Appendix F. The specific structure of any given part

of the measurement system is a function of

the technical requirements of the end product, the legal obligations of the organization, the financial resources available, and

the standards, apparatus, and test equipment

available to meet the ultimate measurement

renui rements

.

2.4.4 Regulatory Agencies

Table 10 gives the titles of regulatory

agencies and agency categories impacting the

Electrical Measurement System.

Table 10. Regulatory agencies and agency

categories

Federal Government

1. Nuclear Regulatory Commission

2. Environmental Protection Agency

3. Federal Aviation Administration

4. Food and Drug Administration

5. National Highway Traffic Safety

Administration

6. Occupational Safety and Health

Administration

State Governments

State Public Utility Commission

The Nuclear Regulatory Commission (MPC)

has the function of being Federal watchdog

over radioactive materials in this country

for both security and safety purposes. They

have issued regulations which compel the formation of quality assurance programs in nu- clear power plants and fuel reprocessing facilities. Regulation 10CFR50, "Licensing of

Production and Utilization Facilities" has

an Appendix B entitled "Quality Assurance

Criteria for Nuclear Power Plants and Fuel

Reprocessing Plants." This Appendix and

corresponding Regulatory Guides call for

periodic calibration of all measuring equipment. The result of such regulations is the

establishment of standards and calibration

laboratories at all nuclear use and processing sites to ensure the quality of measurements at such sites.

The Commission is presently in the process

of revising these regulations to improve

their clarity. The National Bureau of Standards is anticipated to be called upon to provide technical assistance in this activity.

The Environmental Protection Agency (EPA)

has an indirect effect on the Electrical

Measurement System. The instrumentation,

which is required by the agency and by industry to monitor pollution, is used mostly

for analytical and accoustic measurements.

Sales of such instruments are increasing at

a rapid rate, sparked by large increases in federal and private research and development

in this area To], This equipment is basically electronic in nature and, as such, re- quires maintenance, if not electrical calibration.

The Federal Aviation Administration (FAA)

is responsible for all air travel safety in

the U.S. It regulates the communications

and navigation equipment used in the air

travel system. A measurement laboratory facility is maintained in Atlantic City, MJ,

which supports a network of navigation aids

such as radar systems, landing systems, and

beacons at airports throughout the country,

in addition to a test facility for avionics

on the Atlantic City site. In addition, the

FAA is now in the process of implementing a Drogram for the periodic recall and calibration of all test equipment in its system.

This new calibration structure may be ultimately supported by the standards laboratory

of the FAA maintenance and repair depot in Oklahoma City.

The Food and Drug Administration, as was pointed out previously, is expanding into

the regulation of medical devices. Aside

from the proposed regulatory program for new products described in Sec. 2.2.1.1, they are involved in a program to ensure safety

through the proper level of application of

voltage to x-ray tubes. It was found as a result that the voltage dividers used for

measurement of cathode voltages were poorly

calibrated at best and improper x-ray dosages resulted. Bureau of Radiological

Health support resulted in the establishment

of a calibration service for such dividers

at NBS. The National Highway Traffic Safety Administration is responsible for enforcement

of the application of safety standards to automobiles. The group monitors the whole

field of automobile safety testing. Use is

made of Society of Autmotive Engineers'

specifications to detail methods to he followed in the testing and to prescribe re- quirements for quality assurance. Periodic

test and calibration of measurement instrumentation using standards traceable to NBS are features of SAE Standard J21b. Manufacturers are required to abide by its provisions . The Occupational Safety and Health Administration generates job-related safety reg- ulations. Its major impact on the Electrical

51
Measurement System is its requirement that

certain devices be safety-tested in accredited laboratories. That requirement is ex- pected to be the forerunner of many future

requirements for laboratory accreditation

programs. None of the methodologies yet

proposed embodies a valid scheme to ensure

the proper behavior of standards, instruments, and systems on a continuous basis,

either through careful monitoring or any

other means. A real need exists for a technically-valid scheme for accreditation.

State Public Utility Commissions, in addition to regulating the rates charged con- sumers of electric power, regulate the system by which energy transfer is monitored

to determine total user costs. Watthour measurements are used to quantify

energy distributed within the power system

for metering purposes. The electric power

companies are generally required by the

Public Utility Regulatory Commisions in individual states to supply users with watthour meters accurate to within 27. Refunds

are required for overcharges if the meter

regulation is found to be in error — faster

than nominal ~ by more than 27. While power

companies are entitled to recompense from

users should the meter err in their favor,

they generally do not bill such users be- cause of the bad publicity which could result. Because of this situation, power companies typically calibrate new watthour meters with an uncertainty of 0.1-0.2 7. Measurements at this relatively high level of

accuracy also permit new meters to be inspected using sampling techniques, thus

avoiding the large amount of work required

to inspect 1 00% of the incoming lot. Be- cause this practice is common in the power

industry, meter manufacturers must also use strict measurement tolerances.

Installed meters are generally recalibrated throughout the industry. However,

there is no industry-wide established interval between calibrations. This interval

ranges generally from five to fifteen years

and in most instances is done on a sampling

basis with 100? recal ibration only being

performed on the basis of poor results from

an initial group of tests. These measurements are made with the same uncertainty as are those on new meters.

Of all quasi-regulatory activities, procurement regulations governing suppliers of

the Department of Defense and the National

Aeronautics and Space Administration have

the greatest impact on the measurement system. The present hierarchical system of

laboratories, accuracy needs at the primary

standards level, and the generation of use

of detailed calibration procedures all stem from the enforced application of MIL-C45662a , MIL-q-9P59a, MIL-I-4520Ca , and the

equivalent NASA regulations, all detailed

previously.

2.5 Direct Measurement Transactions Matrix

2.5.1 Analysis of Suppliers and Users

The organizational input-output matrix in

Figure f is a portrayal of the present magnitude and importance of measurement tran- sactions among various segments of the economic system. Each entry in the array gives

the magnitude, role of changes, import, adequacy, and type of transaction according to the code in Figure 7. An "X" as a particular entry indicates that the transaction was excluded from the study for some reason. Entries of "0", "X", or "?" require no further information for that transaction. It must be remarked that the matrix is only a qualitative portrayal. The uncertainties

in each ranking factor are high (±2) as it

is difficult to quantify relationships such

as quantity or volume in the context of various sizes and technical contents of industries. The general approach taken was that

a measurement capability could not cease to exist, since in that sense, the "Importance"

entry would be 4 by any rational consideration for each transaction. In addition,

those industries which require measurements,

instrumentation, and standards closest to the state-of-the-art" naturally are more familiar to the authors and received more attention during the course of the study as they are the most likely source of measurement problems.

2.5.2 Highlights re Major Users

Although virtually every segment of man- ufacturing depends to some extent on the

ability to make electrical measurements,

major users of the measurement system have

a very stringent need for an accurate mea- surement capability. These are the aero- space industry, the calibration and repair

industry, civilian agencies of the Federal

government, electronic component manufacturers, computer manufacturers, the De- partment of Defense, the instrumentation

industry, the power industry, and research

institutes. It should be understood that

technical measurement problems to be found

in this system are generally not unique to

a specific segment of industry in the SIC

sense. Rather they are generally associated with a broad class of activity such

as instrument maintenance or quality control of electronic products and accordingly

are not belabored in this section. Pressing

problems which are industry-peculiar are mentioned.

52
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TRANSACTIONS

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3 u 2

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4 Instrumentation

Industry

Rir 1ft 2 BCD

3

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2 CD

2

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3 B-E

3 4

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3 CE

3 f1

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3

3

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2 BCD

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3 31

3 BCD

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6 Other US National Standards Authorities X

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7 State and Local Office of Weights & Mpa«iirp<

X X X

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8 Standards & Testing Laboratories 4 V 1 1

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' Federal Regulatory

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2

1

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3 E X

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2 2

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2

1

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X X X

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13 Industrial Trade Associations

X

3 41

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3

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1 2

2

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1 2

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1 2

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1 „ 2

3 OE '2 23 DE ^ Agriculture, Fishing

Mining

SIC Div. A&B

X X X X X X X X X X X X X X

1

« h 15 Construction

Div. C

X X X X X X X X X X X X X X

16 Fcod, Tobacco, Textile, Apparel , etc

.

SIC =5 20-26:11

X X X X X X X X X X X X X X X

17 Chemical, Petroleum,

Rubber. Plastics etc.

sir ?r-w i?

2

1

1 2 D

3

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18 Metal Products

SIC 33-34, 391

2

1

1

2 D

3

,

1

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2 2

2

2 DE

2

2

2

2 E 1,12 A

3 2

1

2 A

X

1 ,2

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X X X X 2,2

2 B-E

X X X X X x

19 Non-Electrical Machines

SIC 35

2

1

1

2 D

3

1 1

2 E

2

3

2

3 E

2

1

1

2 E

3

3

1

2 A

3 2

1

2 A

X

1 ,

2

3 E X X X X X X X X X X X

20 Electrical a Elec- tronic Equipment

SIC 36

3

3

1

2 0

2

3

2

3 DE

2

2

1

2 D

3 41

2 E

2

3

'

2 E

2

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3

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'

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1

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2

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3 3

1

2 A

3

3

1

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X

1 4

2

! AE

3

2

1

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X K

1—X

Transportation Equip

SIC 37

3 41

2 D

2

3

2

3 DE 2,12 D

3

4

1

2 E

2

4

'

2 E

3 3

2

3 E

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4

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2

3

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2

3

1

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'- 2

1

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;

1 1

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22 Transportation & Publ ic Util ities SIC Div. E

2 21

2 D

2

3 2

3 DE

3

3

1

2 D 1,12 E

2

4

1

3 E

2

3

2

3 E

2 3'

2 E

2

3

3

2 A-D

X

1 22

3 E

X X X X X X X X

3

3

'

3 BE

X X

3 4

1

3 A 22 Trade, Insurance, Finance, Banking etc

SIC Div. F-H h J.I27

X X X X X X X X X X X X I

24 Health Services

SIC 84 X

2

2

1

2 E

3

2 2

2 E

2

1

1

2 E

3

3

2

3 E

3

1 3

2 E 3,13 A

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X X X X X X X X X X X 3

1 2

3 B-E

25 General Public X

1

1

1

3 E X X X X X X X X X X X » [3 V

Figure 6. MATRIX

53
B C

A

D E

A. Volume or quantity of transaction

0 - Trivial (other entries not needed)

1 - Minor

2 - Moderate

3 - Above average

4 - Major

X - Excluded from study

B. Importance

1. Convenience; not essential

2. Economically important

3. Legally required; no alternative source

4. Absolutely essential

C. Adequacy of Transaction

1 . Under control

2. Marginally adequate

3. Deficient, but useful

4. Out of control; totally inadequate

D. Rate of change of transactions

1. Declining volume (negative)

2. Stable (0)

3. Growing (+)

4. Explosive growth (very +)

E. Type of transaction

A. End use measurement data (specifications, etc)

B. Other measurement services (calibrations, etc.)

C. Measurement instrumentation and software

D. Measurement "How to" information

E. Measurement requirements information

F. Legal performance specifications

Figure 7. Code for transaction matrix

54
Aerospace (SIC Codes 3721 , 3724, 3728, 3 761 , 3764 , 3769, and 3662). The "aerospace industry is that segment of manufacturing which

produces space vehicles and aircraft and

their supporting electronics systems for

both military and civilian applications. It is a high technology industry (engineers and

scientists employed, 161,000) which makes ex- tensive use of electronics in the develop

ment and manufacture of its products, and es- pecially, for their weapons and guidance

systems. The industry is a large one, with

estimated sales of products and services in 1974 of $20.9 billion of aircraft and parts,

$9.6 billion of electronic systems support

equipment, and $5.8 billion of guided missiles and space vehicles [9]. Estimated total sales (1976) are $35.7 billion with an estimated electronics content in excess of

$20 billion. The estimated exports value of

aerospace equipment for 1976 is ^7.6 billion,

the third consecutive year that this industry's exports have exceeded $7 billion. This

industry embodies 2,107 establishments and

employs about 928,000 workers [10].

The primary impact of this industry is upon our national defense. Slightly over

ten percent of the aircraft produced were

sold to the Department of Defense as were

all guided missiles. These products are vital to the national security. Important sec- ondary benefits have come from this source. Consumer electronic products have been revo- lutionized by such spin-offs from the aerospace industry as monolithic large-scale integrated circuits whose low cost, reliability, and small size per unit function have

permitted the production of quartz crystal

watches, pocket calculators, and miniature

radio sets at exceedingly low cost.

An increasing portion of the output of

the electronic systems component (SIC Code

3662) of this industry is being consumed by

civilian-oriented commercial and industrial

firms. This is due to the need in industry

to make more use of digital data transmission, automated process and quality control,

and telecommunications to offset rising labor

costs and lower profit margins. Such areas

as satellite communications systems, data net- works, land-mobile radio communications, and

interactive, broadband cable communications

systems contribute heavily to this increase.

The electronic systems component is an area in which exports are expected to rise

sharply, 23%, from 1975 to 1976. This is largely due to increased consumption in the

developing nations and, particularly, the

oil-producing nations of the riddle East. These countries, particularly Iran, Saudi

Arabia, and Indonesia, are attempting to develop their industrial bases in the avionics,

industrial, electronics, and communications

areas, in addition to increasing their military sophistication [101. Exports in this area are expected to increase for the next

five years.

Electrical measurements in the aerospace

industry are used to characterize electronic

circuitry of all types, both for the assurance of quality and compatibility in manufacture and for maintenance while in service.

Accuracy requirements for so-called end mea- surements are ouite high (as high as 10 ppm,

in some cases) due to the sophistication and

precision of guidance and weapons systems.

Since nearly all companies in this category

are contractors or subcontractors to the De- partment of Defense or to the National Aeronautics and Space Administration, the industry is subject to the provisions of the regulations used by these government agencies to assure quality in the systems and vehicles

procured. The concepts there outlined are instrumental in making the accuracies re- quired of an individual corporate calibration

support structure quite high.

Measurement problems faced by the aero- space industry are as follows:

(a) to reduce the cost of quality control

efforts in the face of a slightly deflated

market while adhearing to DoD and NASA re- quirements and regulations [10,13,15]; and

(b) measurement problems inherent in the

use and support of automatic test equipment,

such as those resulting from failure to calibrate under use conditions (e.g., making dynamic use of A-D converters which have been

calibrated statically) [15,16].

Calibration and repair of test equipment

(SIC Code 7393) . There is a small industry

in the U.S. having to do solely with the

calibration and repair of instruments, test equipment, and standards. Most companies

involved in this activity are quite small,

having annual incomes of less than $500,000

and employing less than 25 people. These

companies supply calibration and repair ser- vices to a variety of industries, such as small component manufacturers, transducer

makers, small electronic job shops, and

other companies whose calibration needs are

so modest that the establishment of an internal capability would be unduly costly.

(Such service is provided also on a fee basis by the corporate standards laboratories

of many large corporations. See Appendix D. ) Equipment used by such companies ranges

from state-of-the-art standards and calibration equipment to older test equipment of

relatively poor accuracy. This results in a wide range of capabilities being available,

from the state-of-the-art to meter accuracies, that is, three to five percent. In

55
most cases the capability is equal to the

job performed. Problems related to electrical measurements in this industry seem to be

logistical in nature. Lack of facilities

for training, poor communications of technical data, increasing costs of publications,

and an increasing shortage of experienced

technicians are a few examples.

Civilian agencies of the Federal government

(SIC Div. 1). Those civilian agencies which

have or support (as opposed to require) an electrical measurements capability are as follows: Federal Aviation Administration,

Bureau of Reclamation, Energy Research and

Development Administration, National Aeronautics and Space Administration, Tennessee

Valley Authority, and National Oceanic and

Atmospheric Administration. FAA activity

was covered in Sec. 2.4.4. The Bureau of

Reclamation operates Hoover Dam and leases

power generated thereby to five regional

power companies. The organization has a standards laboratory that supports all of

the maintenance and power monitoring equipment on the site.

The Department of Energy (DOE) has a con- tracted primary standards laboratory in Albuquerque, NM, run by the Sandia Corporation, which is responsible for supporting and

assuring quality of measurements and contractual compliance with calibration requireemnts at each of the DOE field sites. The

National Aeronautics and Space Administration

(NASA) maintains a large number of standards

laboratories, one at each of its major centers and tracking facilities. NASA has an extremely large test equipment inventory

which is used to support space electronics,

its computer network, communications equipment, tracking systems, and aircraft. Many

of these laboratoies are staffed by contractors but the agency owns all of the equipment.The Tennessee Valley Authority laboratory

at Chickamauga Dam, TN, supports the power

monitoring equipment and electronic maintenance areas at that facility. The National

Oceanic and Atmospheric Administration supports the test and measuring equipment it uses in its investigation of undersea and

atmospheric phenomena by maintaining a standards and calibration facility at the Navy

Yard Annex in Washington, DC. The social impacts of these agencies are wide-ranging. The Department of Energy

plays an ever more important role in the

search for means of alleviating our energy

shortage. Since it is a highly technical en- deavor, the use of measuring apparatus is heavy, and electrical measurements, especially those at high voltages, are of prime

importance. NASA is concerned not only with

space ventures but also with air transportation problems. Technical spin-offs of its

programs have resulted in the creation of

whole new technologies such as integrated

circuit technology.

Electronic components (SIC Code 3 f 7 )

.

The

industry employs about 359,000 people in about 2,500 establishments. The industry's

gross sales were $10 billion in calendar

year 1975, and are projected to exceed $11

billion in ]°Jf [10]. Fxports in this field

are at the $1 billion mark and increasing

rapidly; up 23" from 1974 to 1975. It produces such components of electronic equipment as resistors, capacitors, inductors,

diodes, transistors, integrated circuits,

tubes, and rheostats. By doing so, the industry provides versatility and cost effectiveness for every electronics-intensive industry such as those involving the production of automotive electronics, communications equipment, instrumentation and process

control equipment, etc. Measurements are vital to this industry

for both product duality control and development of new products. The biggest measurement problems faced here are those concerned

with the proper calibration of very highspeed component testers. Such equipment is used to test large numbers of resistors, ca- pacitors, or other components at a high rate

of test. Even more importantly, complex integrated circuits must be thoroughly tested

in a short time frame. The time taken per

measurement is generally much shorter than

the time per measurement allowed in the calibration process, and for that reason the

calibration data may not be valid when the

tester is used on the production line.

There are a few problems having to do

with the state-of-the-art in measurements.

For example, solid-state device manufacturers are generally convinced that some of the

voltage reference devices they produce have

stabilities in the sub part-per-mi 1 1 ion (ppm)

range over the short-term and long-term (a few weeks) may be as stable as 1 or 2 ppm.

Few, if any, of these same manufacturers,

however, have a measurement capability adequate to determine this. Such capability

would mean a cost savings to industry as a whole because if these devices are indeed

capable of such stabilities, more accurate

instruments could be built and instruments

being built at present accuracies would re- quire less frequent calibrations and possibly less maintenance. Even if the refer- ence devices were not found to be as stable,

an improved measurement capability would re- sult in the ability to determine their sta- bility with fewer measurements taken over a shorter time span. Manufactured devices

56
could thus be much more easily categorized

and sorted. Similar problems exist in the

scaling processes for other units such as resistors.

Computer manufacturers (SIC Code 3 573).

Sales of the computer industry were $10.4

billion for 1975 and are estimated to in- crease to $12.1 billion in 1976 Mo]. This

rapidly growing industry (current growth

rate is about 14' per year) is estimated to have 179,000 employees. Its output products,

computers, large and small, and peripheral

equipment, are being used more and more throughout our society. They are used as ac- counting machines, problem solvers, process

controllers, navigation aids, machinery con- trollers, and in nearly every other situation where decisions must be made and action

taken on the basis of numerical data. The

advent and continuing improvement of the

iPinicomputer, a small, inexpensive, and re- liable computer, and the microprocessor, a computer on a chip, have made the tremendous

power and capability of the computer available to virtually every segment of industry.

Small scale processes are being automated at

a very rapid rate, thus providing some offset to the ever-rising cost of labor in this

country.

Since most computers are electronic, electrical measurements play an important role

in their manufacture. Each component that

is received is checked to its specification

at most companies. Testing is done at each

stage of the assembly of such a device. The

equipment used for all such testing must be

maintained and calibrated and most large com- puter companies have corporate standards and

calibration laboratories to perform this

function. Electrical measurement problems

in the computer industry are related to the

use and support of the automatic test sets which perform the quality assurance function.

Some problems result from applying static

calibration data to dynamic testing situations. A particular problem of great current interest results from the giant number

of measurements required to test a digital

circuit to an adequate degree. Even if it

is solved, however, there will be increasing

need, as semiconductor components become

faster responding and more complex, to make

measurements of both digital signal response

and dielectric properties of components at higher speeds and with greater reliability.

The state of technology is pushing the few

standards which do exist in this area. An- other problem area is the measurement of imDedances, especially inductors, in the frequency range between 10 kHz and 1 MHz.

The Department of Defense. The Department

of Defense is very heavily involved with

electronics in communications, weapons, guidance, and navigational systems. The Depart

ment will have spent S2.6 billion for re- search, development, testing, and evaluation

of such eauipment and Si. 61 billion for the

procurement of electronics systems for FY 75 Total spending for FY 75 on electronics prod

ucts as estimated by the Department of De- fense is S9.fi billion [9,10]. Assuming a five-year lifetime of electronics equipment

in the military, using marketing data for

the past five years such as surveys performed annually by Electronics magazine, one

can calculate a Sfi billion inventory of

equipment which must be maintained. This is done with a huge inventory of test equipment

supported by a hierarchical laboratory sys- tem in each of the four services. To illustrate the magnitude of the problem, the 1972

edition of Air Force Technical Order

33K-1-1PO lists by part number over 28,500

individual items in the inventory requiring

calibration periodically. In excess of sixty percent of this inventory requires electrical measurements p7] for its support.

The Department of Defense is the largest

single consumer in the country and, accordinaly, wields a trememdous impact on the mar

ketplace. Technical spin-offs from its programs, much as is the case with NASA - who,

for example, are credited for the accelerated development of large-scale integrated

circuit technology - have served to advance

the technology base of our society.

The Department maintains a system of inspectors in various organizations (Defense

Contract Administration Service; Air Force,

Navy, Army Plant Representative Offices;

etc.) to ensure compliance with the calibration and quality assurance providions of the

procurement specifications listed in Sec. 2.2.1.1. In this way, it influences the na- ture of the measurement system in a large

segment of industry.

As mentioned before, the main use of elec

trical measurements within the Department of

Defense is to maintain its extremely large

inventory of electronic systems and equipment and the test equipment used to support

it. The major current problem areas are:

(a) the support of automatic test and

checkout eauipment ['13,16];

(b) the support of eauipment at remote

locations; and

(c) in the face of decreased defense

spending, to optimize the entire calibration and maintenance structure |"13,15].

Problems in each of these areas have two aspects. The first is the need for dynamic

standards and methods for their use to support or verify measurements made under time

57
constraint with automatic test equipment, es- pecially that under computer control. Such

standards simply do not exist and their ab- sence both prevents direct verification of

the quality of individual measurements made

by present equipment and limits the advance

of automatic testing by confining its application to the measurement of those parameters for which static standards exist. For example, the so-called third generation ATE system makes measurements using sampling

techniques and arrives at final results

through harmonic or Fourier analysis software. System capabilities are given, how- ever, only in terms of derated rms or average voltage measurement specifications as lack of knowledge of the information content

of a single sample and a lack of arbitrary

wave form standards precludes the precise

calibration of such a system. (Specifications are derated relative to the potential

accuracy of the sampling device.)

The second aspect has to do with the

achievement of adequate measurement accuracy

at acceptable cost. Accuracy requirements

for the support of end items in the military

inventory are gradually becoming more stringent as electronic components and systems

increase in quality and sophistication and

improved reliability is sought for strategic

or economic reasons. At the same time budget decreases result in reduced resources

for measurement support and quality activities. Support equipment for new systems

and test equipment used in development and

repair activities include new instruments

with performance characteristics rivaling

the best measurement capabilities of the

typical DoD calibration laboratory.

These factors result in the need for the

development of new calibration procedures,

better training of technicians and, in some instances, exhancement of calibration and

measurement accuracies through new standards

or more sophisticated measurement experiments. Automated systems for the calibration of general purpose test equipment, the use of MAP techniques for the support of

line calibration laboratories, and in situ

tests of both ATE systems and electronics

in aircraft, field installations and surface vessels are all being employed to deal

with these problems.

Detailed technical problems within the

system are dealt with by the professional

staffs of the Air Force, Army, and Navy metrology engineering centers. Specific, long- term problems are dealt with by these people.

In those instances where the advancement of

measurement science or new standards are re- quired, the DoD enlists the help of NBS

through the sponsorship of engineering projects. Typical objectives of such projects

are the development of an al 1 -cryogenic voltage standard accurate at the part in 1 f)8 level; the design of a low-frequency, lowvoltage measurement instrument to enhance vibration measurement capabilities; and the

development of an automated capacitance

bridge with which DoD primary laboratories

can provide improved service at reduced cost.

In addition to providing this measurement

science support, the NBS serves as a basis of

measurement uniformity for the DoD system by

supportinq the four primary standards laboratories through calibration services and MAP activities.

The instrumentation industry. This industry

was covered in Sec. 2.2.2.2.

Electric power industry (SIC 3825, 3511, 361,

362, and 49 1 . The electric power industry

has two components, the generation and distribution of power (SIC 491) and the manufacture of equipment for the generation and mon- itoring of electric power.

The generation, distribution, and sale of

electric energy is the largest single business in the U.S. Approximate sales in 1973 were $31.7 billion and in 1974 were $39.1 billion. In 1975 the industry sold over 1.7

trillion kilowatthours to nearly 82 million

customers for an estimated revenue of $47.9

billion [18,19].

The production of turbines and turbine

generator sets produced shipments valued at

$2.4 billion in 1975. Shipments for 1976 are expected to increase by about eight percent

to $2.5 billion. Seventy-three businesses

employ 47,Onn workers in the production of

this equipment T18]. Production of transformers, switch gear, and industrial controls for

power distribution (SIC Codes 3612, 3613,

3622) totaled $6.7 billion in 1975 and is ex- pected to reach $7.1 billion in 1976. More

than 200 firms are engaged in the production

of transformers alone and employ over 50,000

people r Q J0]. ETectrical measurements are required by this seqment of the industry for

quality control purposes. The levels of ac- curacy to be found are generally lower than

those at the state-of-the-art and readily

achievable with off-the-shelf instrumentation, but they are required to enable quality of these goods. No major metrology

problems are evident here.

In addition to being one of the key industries in the U.S., the electric power industry supplies most of the rest of industry

with motive energy. This is done through a system of interlocking grids of independent

power companies. Electrical measurements

are necessary for the monitoring and control of the power flow in the qrid as well

58
as for energy metering by both power companies and their customers. Measurement problems include:

(a) the determination of transformer

ratios at high voltages;

(b) the characterization of high voltage

capaci tors

;

(c) the measurement of the dissipation

factor of high voltage capacitors;

(d) the accurate calibration of watthour

meters

;

(e) the development of techniques for

measurements in support of research to improve the transmission and distribution of

electric power;

(f) the measurement of transient signals

at high voltages; and

(g) the development of time-of-day metering techniques to enable the use of a rate scale in which the price of electricity in- creases during periods of peak demand.

NBS is involved in each of the above

areas. The High Voltage Measurements Section is engaged in research and development

programs generally supported by the Electric

Power Research Institute and the Department

of Energy designed to achieve the measurement capabilities at high voltages needed to support both basic research and power system

operations. Examples of such projects are those to provide capability for measurement

of pulses and transients of up to several

million volts, to develop systems for the in

ditu calibration of coupling capacitor voltage transformers at voltages up to megavolt

levels, and to improve the standards used

for the measurement of current and power at power line frequencies.

In addition, NBS calibration services

provide the basis for uniform measurement

of electrical parameters necessary to dayto-day power company operations. NBS calibrations of standard cells, resistors, and

watthour meters stand behind all measurements made for metering purposes, including

those using the watthour meters found on every home in the U.S. Companies rely on

NBS calibrations of high voltage capacitors,

transformers and shunt reactors to maintain

the efficiency of the power distribution sys- tem throughout the country. These services

could not be provided without a strong, on- going research effort in high voltage measurement technology such as exists in NBS

today

.

Research. In this category fall all independent research organizations, universities,

and national laboratories. Since much of

the research done in every field is experimental, the ability to make accurate electrical measurements is a necessity. An example

of how precise measurements are needed is seen in a program at the University of Penn- sylvania for the study of the electrical

properties of solid-state, cryogenic

Josephson junctions. The purpose of this

work was to determine an experimental value

for the quantity e/h with sufficient accuracy to enable its use in the verification of

quantum el ectrodynamic theory. To do this,

a measurement of its voltage output had to be made with the maximum available accuracy,

that is 0.1 ppm relative to the legal volt.

This necessitated the construction of a special instrumentation and the initiation of a special effort, both at NBS and the University of Pennsylvania, to verify the accuracy

of the University's voltage standards.

A second example is to be found at Argonne National Laboratory, where extremely

precise calorimetry work requires that voltage and resistance measurements be made with

accuracies in the part per million realm.

The experimenter is compelled to keep his own set of standards which are calibrated at

NBS.The energy crisis and the resulting perception that electricity will become the dominant means of energy utilization has fostered research, sponsored primarily by the De- partment of Energy (DOE) and the Electric

Power and Research Institute (EPRI), for the

increased generation and improved transmission of electric power. This research is being carried out in industry, at universities, and in various national laboratories,

such as Brookhaven and Oak Ridge. Typical

projects are the investigation of the use of

superconducting transmission lines, ultrahigh voltage overhead lines, direct current

transmission, solar energy, and cryogenic

motors and Generators. Approximately $14.3

million has been invested in FY 76 for trans- mission and distribution research by DOE and

52^. f million by EPRI for calendar year

197P fl9].

The success of such research hinges on the ability to make accurate measurements at high voltages and under difficult environmental conditions and in many cases adequate

techniques and standards for such measurements do not exist. As mentioned in the previous section, NBS is undertaking a number of

research projects to establish the reauired

measurement methodology. Examples of projects specifically bearing on energy-related

research are those of applying the Kerr effect to map electric fields, the design and

construction of pulsed current and voltage

generators and associated measurement apparatus to enable calorimetric determinations

of thermal /physi cal properties of materials

used in nuclear reactors, and the development of calibration techniques in support

of instrumentation used to determine the

59
effects of fields associated with voltage

transmission lines on plant and animal life.

In other areas, NBS work in the fundamental constants field provides data contributing to the verification of physical theory,

such as quantum el ectrodynamic theory. MBS

has ongoing efforts in areas requiring precise electrical measurements. Determinations of the ampere, volt, Faraday, ohm, and

the gyromagnetic ratio of the proton are presently underway. NBS has participated in the most recent (1973) least-squares adjustment of fundamental physical constants published by CODATA, the Committee on Data for

Science and Technology. Another such adjustment is to be carried out in 1980.

As has been shown, the National Electrical Measurement System pervades most of the

major industries and government agencies to

a larger extent. As might be expected, mea- surement problems and requirements stem from

two sources, the use of electronics as an implementor of manufacturing quality and

process control, computations, and communications and the electric power industry and its

needs for efficient power generation, distribution, and metering. Measurement technology

in the former instance covers the ranae from

dc to 1 megahertz at modest levels of voltage and power whereas in the second case mea- surements are generally of electrical parameters at dc and 60 Hz and high power and

voltage levels.

Through its calibration and research ac- tivities, NBS provides the measurement sys- tem with its base, thereby ensuring needed

stability and accuracy in terms of SI units.

High-technology industries with the most

stringent accuracy requirements provide the

interface with NBS needed to enable virtually all electrical measurements made in the

U.S. to be expressed in terms of the national

units. Measurement accuracy in areas of more

than modest requirement is ensured by systems

in which the instruments used to make the

measurements are calibrated periodically us- ing standards or test equipment with accuracy

capabilities from four to ten times better

in each case. A chain composed of such calibration links reaches from most commercial instrumentation back to standards supported by

NBS. The system remains viable due to mea- surement research performed at NBS and in the

instrumentation industry.

60
3. IMPACT, STATUS, AND TRENDS

OF THE MEASUREMENT SYSTEM

3.1.1 Functional, Technological, and

Scientific Applications

The use of electrical measurements generally falls into one of four categroies - the

production of reference data, setting and

testing to specifications, maintenance and

reliability activities, or the control of

manufacturing processes. Reference data, including physical and engineering data covered

in Section 2.2.3, is vital to designers of

virtually all high-technology products. The

Electrical Measurement System ensures that

such data have universal meaning by providing common reference bases in the electrical

units, both for those experimentalists who

produce the data and for those designers who

must use the characteristics of materials,

circuits, etc. in an engineering application.

The next major use of electrical measurements is in vendor determination of performance characteristics of manufactured products and customer testing of those same devices to ensure that their specifications or

the manufacturers claims are actually being

met. Most commercial electronics and electrical merchandise is traded in this fashion.

Electronic components, modules, instruments,

devices and systems are generally tested,

sometimes on a sampling basis, to specifications prior to their acceptance. The measurement system is the foundation for all aspects of this type of activity. The

availability of universal units of measure

permits interchangeability of electronic

components and modules (a must for the mass production and subsequent maintenance of complex electronics) and multiple manufacturers

of commonly used circuit elements. These

factors, coupled with inexpensive production

techniques, enable all electronics products

to be produced inexpensively and to function

with a high degree of reliability.

Electrical measurements are vital to maintenance and reliability functions throughout

the electronics user community. Test equipment is used both to ensure the proper operation of functional electronic equipment, such

as radar sets, inertial guidance units, etc. used in critical applications (i.e., space

and weapons systems) on a periodic basis and

as a means of trouble-shooting and repairing

nonfunctional electrical and electronic equipment. This test equipment is generally

checked or calibrated periodically using more

accurate standards and equipment through

the echelon system of standards and calibration laboratories which exists throughout industry and most government agencies involved

with technology.

The final major use of electrical measurements is for the control of industrial manu- facturing processes. Early automatic control of manufacturing processes was achieved

through pneumatics and electro-mechanical

servomechanisms. Besides being a prime source of power for the process itself, pneumatics gave the design engineer the ability

to implement various control algorithms tending to optimize the process. The advent of

electronic amplifiers enabled the implementation of these same functions electrically

with many control adjustments difficult or impossible to achieve pneumatically. Also over the years the development of transducers,

devices which produce electrical signals of

a known relationship to the magnitude of a particular physical quantity, has gradually

extended the versatility of electronic control. The advent of the inexpensive digital

computer along with electronic instrumentation with digital data outputs has hastened

the application of electronics in the control

field. Control systems based on digital com- puters have the advantages of speed of res- ponse, economy, and software signal conditioning. Not only does the latter make possible the use of exceedingly complex control

algorithims, but it also permits extreme

flexibility of modification without extensive hardware changes.

The most stringent measurement requirements come from the instrumentation industry,

the aerospace industry, and the Department

of Defense. The DoD and the NASA generate

stringent requirements for the accuracy and

reliability of guidance, weapons, and navigation systems. Accuracy requirements of

better then 0.001% for various electrical

parameters exist in some of the new operational equipment itself. This level of re- quirement places a burden on the instrumentation industry which is called upon to provide the test equipment and support stan- dards necessary to the development, production, testing, and maintenance of the operational equipment. Further the test equipment

and standards themselves must he verified and

maintained in all three of these areas of

the economy. NBS plays a vital role in this

function by providing calibrations of stan- dards, Measurement Assurance Program ser- vices, reference data, and training services.

Less accurate — but in some cases no less

stri ngent — requi rements come from the electric power and the semiconductor industries.

In addition to the routine measurements of

electric power for revenue purposes, the

power industry reouires many difficult mea- surements under field conditions for both

operational and research purposes. Examples

of these are given in the previous section.

Large scale integration techniques have per61
mitted a great growth in complexity of circuit elements. Meanwhile, the ever-decreasing cost of these same products has lead to their use in many applications in which re- liability is critical, such as electronic

control of automobile brakes, medical life

support systems and instrumentation, and aircraft navigation systems. Testing of these

circuit elements must be done, therefore,

accurately and rapidly to achieve utility

and low cost. This simple factor has lead

to a surge of interest in automated testing

and requirements for support of electrical

measurements made "on the fly". There is a plethora of dynamic data conversion devices

on the market with which such measurements

may be made. Lacking are the standards

which form the means by which a translation

may be made from traditional standards and

measurements, which involve no time con- straints, to the new requirements for dynamic measurements.

On the other end of the measurement spectrum lie those electrical measurements which

are part of the day-to-day experience of the

typical man on the street. Examples are those made in the course of tuning an automobile; for repairing television, radio, or high-fidelity audio equipment; to maintain

home consumption of electric power; and to provide for the safe navigation of commercial airlines. Instruments used in the for- mer two cases are generally only supported by

the measurement system upon their failure.

The accuracies required in support of most consumer services are readily obtainable

since they lie in the 1 to 5 percent range.

Electrical measurements are used to control the quality of any item manufactured

by an automatic process. The use of transducers, which convert other physical parameters into electrical parameters, coupled

with electrical instruments with digital outputs enable any physical or chemical production process to be run under computer control. Thermocouples, strain gages, tachometers, and fluid level meters are examples

of transducers. Table 11 lists some articles whose production is made possible by

electrical measurements.

3.1.2 Economic Irpacts - Costs anH Benefits

The costs incurred by the economy annually

for electrical measurements are very low by

comparison with benefits received. A survey

of the manpower available in the calibration

laboratories listed in the NCSL Directory of

Standards Laboratories shows a total of P78

engineers and 320P technicians for all mea- surement areas [14]. A conservative estimate

was made that there are roughly 1000 measurement engineers and 5000 technicians in that

Table 11. Representative heavily

measurement-dependent products

Electronic

Components (resistors, capacitors, etc)

Computers

Digital watches

Electronic calculators

Frequency standards

Guidance systems

Navigation equipment

Stereo receivers

Other

Engine blocks

Oil , gasol ine

Plastics

Rayon, nylon fabric

Steel

Synthetic rubber

portion of the measurement system which provides standards and calibration support to the remainder of the system. Approximate

labor and overhead costs for that number of

people are estimated to total no more than

$130 million per year. The remainder of the

system is many hundred times larger and difficult to quantify. The degree of capital

i ntensi veness of the measurement system is illustrated by the annual expenditure of over $5.6 billion for new measurement instrumentation, of which $1.3 billion is test equipment r9l.

The above manpower figures are totally

exclusive of test engineers and technicians

whose functions are not primarily concerned

with calibration measurements. Virtually

every electronics engineer (except digital

logic designers) must rely on electrical

measurements as a tool necessary to the

performance of his function.

Over the past ten years, in excess of . $9d billion worth of electronic systems and

communications equipment were sold in the

U.S. riO, 20]. All of this eouipment re- quires maintenance support, and much re- quires periodic calibration to insure its

proper operation. During the same period

over $11 billion worth of electronic test equipment was sold for that purpose [9,10].

All of this apparatus requires periodic

calibration. The scientific apparatus inventory is estimated at $12 billion for that same 10-year period [9].

The electronics market as a whole is supported by this measurement capability. Gross

sales for the industry were forecast at $64.9

billion in 1977 [21]. New automated systems

developed explicitly for testing integrated

62
circuits, which are becoming more and more

complex, enabled achievement of this sales

figure. This represents about a

9°' growth,

despite materials shortages and a aeneral

economic decline.

3.1.3 Social and Human Impacts

The largest social impacts of the electrical measurements system lie in the medical

field and the public safety areas. Modern

instrumentation has made possible great

strides in diagnostic medicine. The bodies

of patients in intensive care units are monitored on a twenty-four hour basis by com- puter-controlled instrumentation systems.

These systems not only alert medical staff

members to impending crises, but provide a powerful tool in the analysis of the pa- tient's condition. There are under development automated systems which would give physical examinations to individuals without

requiring a doctor in attendance. Electronic

measurement equipment and computers form the

basis for new systems to perform internal

scans via ultrasonics or x-rays.

In the safety area, measurements are vital to ensure the reliability of electronic

equipment such as braking and ignition systems for automobiles and navigation and landing systems for aircraft. The failure of any

of these systems could be calamitous, whether

due to component failure or electrical interference. Reliable measurements are necessary

to avoid failures from either cause.

3.2 Status and Trends of the System

The electronics sector. At the present

time the National Electrical Measurement System is capable of handling nearly all of the

traditional measurement needs of our economy.

The accuracies available within the system

for so-called static electrical measurements,

i.e., those in which the measurement is not constrained to take place within a particular

time interval, far exceed any technical accuracy requirements. The common practice of

insisting upon accuracy ratios in calibrations of between three- and ten-to-one, a practice fostered by the Department of De- fense and NASA, generally ensures that specified accuracies are delivered within the

calibration traceability chain. However, the

system does have a serious fault which is not entirely technical in nature and which is being increasingly recognized and dealt with.

This fault is the lack of redundancy or feedback in the rigidly stylized, hierarchical

cal ibration/traceabi 1 i ty structure to ensure

that the applied measurements supported by

the system are in fact appropriate and adequate and their quality has not been adversely affected by such factors as environment,

electromagnetic interference, or poor power

lines. Commercial and government organizations are slowly recognizing the need for

periodic sampling of measurements being made

on the production line, in the aircraft, or

at the power substation.

There are several areas in which traditional measurements needs are, at present,

unfulfilled. The most urgent of these are:

(a) characterization and specification

of power line properties. There have been

failures and erroneous data from instrumentation caused by spikes or hash in the power

line. The medical profession and medical

instrumentation industry are most severely

affected by this problem due to the seriousness of potential consequences. A committee

dealing with the problem has been formed un- der the auspices of the National Committee

for Clinical Laboratory Standards. The es- tablishment of specifications and measurement

techniques for interfering signals on power

lines will enable the design of better instruments and provide a basis for settling

the issue of legal liability for consequences of error. This issue arises from

the fact that instrument makers do not specify the limitations of their equipment in that regard and hospitals are not legally

bound to ensure a specified purity of content of the power mains.

(b) measurement of ac voltages at low

amplitudes (<50 mV) and frequencies (<50 Hz).

This measurement capability would be used

with transducers to make measurements of low

frequency vibrations such as those characteristic of trucks and helicopters. The De- partment of Defense and the transportation

industry require the ability to test the sus- ceptibility of various items of critical na- ture to damage by truck or air transportation.

(c) impedance measurements in the upper

ends of the low radio frequency range (20

kHz to 1 MHz). These impedances measurements are needed to support an increase in carrier frequencies in some communications

systems. Increases in carrier frequency

will result in nroportional increases in

message capability.

(d) studies relating to reliability and

the predicition of long-term performance of

components and assemblies based on shortterm, high-precision data needed by the electronics industry for general improvements

in thei r product 1 ine.

(e) measurements of phase angle in the

frequency range from 1 Hz to 30 kHz. Phase

angle measurements are important to the testing of components, solid state circuits, and test equipment used in and to support com- munications and weapons systems. In the lat63
ter, some of the more precise measurements

are required for testing synchro's and re- solvers used to position the antenna of

search and target acquisition radar.

There are two other perceived problems

which affect the system. Technical information is disseminated primarily by word-ofmouth (between colleagues), through sales

efforts of instrument companies, and by technical publications in such journals as IEEE Transactions on Instrumentation and Measurement , Instrumentation Technology , Metrologia , and Measurements and Data . Articles in the

above tend to be academic in nature. De- spite the existence of the Government Industry Data Exchange Program, which has a library of calibration procedures, and the

above-mentioned means of communication, useful technical information on a how-to-do-it

level does not reach the bench-level metrologists and those outside the precise measurements field (i.e., quality control or automatic test equipment engineers). In many instances there is conflict instead of cooperation fostered by ignorance and no means of

communication. NBS is planning a series of

publ i Nations and seminars to help meet this

need.

The second is in the area of manpower.

Many of the highly-skilled technicians and

engineers in the field are on the verge of

retirement. New people in the field tend

to remain there only temporarily. Metrology

may in the future be caught in a three-way

trap between the lack of communications and

informal education opportunities, the decline of the armed forces as a source of ex- perienced people, and the increasing sophistication of new instrumentation, which

requires a higher level of education and appreciation for its proper support.

Two major factors presently affect the

system and will apparently do so for the

foreseeable future. These are the revolution taking place in the electronics industry, and the energy crisis. Advances in

semiconductor fabrication technology and the

application of new semiconductor structures

to circuit design are resulting in the availability of very complex digital and analog

circuits in small, component-sized packages.

As a result an increasing number of standard

circuits, such as amplifiers, computer processing units, power supplies, digital multipliers, analog-to-digital converters, etc.,

are as available to the designer as resistors, transistors, and tubes have been in

the past. Because of this ready availability, electronic equipment has increased —

and will continue to increase - in functional

complexity and sophistication, while its cost has dropped.

The solid state revolution has fostered

a parallel phenomenon in the computer field.

Minicomputers, which have only readily been

available for about the past 10 years, have

so improved that new models offer sophistication which could be found only on large

systems costing hundreds of thousands of

dollars not long ago. Yet a system with a fair-sized memory, reasonable software, and

a mass-storage peripheral can be purchased

for well under $15,000 at the present time.

And, now available are microprocessors, central processing units on a chip, with fairly

comprehensive capabilities. These devices

are inexpensive enough to permit their inclusion in electronic equipment to replace the

hand-wired logic circuitry normally used

there for controlling operations and trans- mitting data.

The above considerations have had and

will continue to have a major impact on both

measurement requirements and the tools or means by which measurements can be made.

The increasing complexity of components,

sub-assemblies, equipment items, and electronics systems have resulted in increasingly

complex tests necessary to ensure quality in the manufacture of such equipment, proper

performance while in use, and affordable

maintenance for it in the event of malfunction. Typically, tests may be comprised of

hundreds or thousands of measurements.

The only way that such tests can be performed in a reasonable, time and with cost puter to control the measurements, acquire

and process the data, and analyse the re- sults. Cost effectiveness is increased by

running the system at a higher rate of speed.

Therefore, the thrusts of the instrumentation industry are in the direction of increased accuracy of measurement at increasing speeds, and of making the instruments

as compatible as possible with computers.

The impact of this industry on the measurement system is heightened by the fact that

the designers of automatic test equipment

(ATE) systems for both military and commercial applications design these systems using

the specifications for off-the-shelf test instruments as building blocks in the design.

This leads to performance problems if the

instruments used do not perform as expected

because of environmental noise, or ground

problems. It may also lead to calibration

support problems requiring high-grade portable standards.

The future will bring even more complexity. The Department of Defense now requires

testability. Many new prime systems will

have the functional capability to test themselves. Most circuits will be designed with

testability in mind, that is, with special

input and output lines with the sole function of expediting testing. Some digital

64
devices will actually contain two similar

circuits with auxiliary circuits to compare

their behavior. Much is being done to lay

the foundation for the development of sys- tem testers, portable devices which can exercise a test system under use conditions

by simulating a typical units under test [13,15,16].

The industrial and commercial sector tends

to have less stringent accuracy requirements

for electrical measurements than the military/aerospace sector. This results from the

lower technology level found in general industry. On the other hand, measurements are required to be made under severe ambient con- ditions. Because of this factor and the need

for a means of coping with a general decline

in productivity, the use of automatic test

and control systems will increase rapidly in this sector over the foreseeable future.

Much effort is being expended in development

of computer-aided manufacturing (CAM) technology, the use of robots, and application

of computer-aided design (CAD) techniques in various areas of industry, as well as in the use of automatic test equipment and transducers for quality assurance. Automatic test equipment as used in the

commercial sector differs in another way

from that developed for the military/aerospace complex. Military and space operations

have a critical nature due to the high potential for loss of life, property, and national

existence. This fact and the often unique

nature of the equipment employed in those

programs leads to the procurement of specially-designed or "custom" measurement instruments and automatic test systems suited for

testing of a specific prime system. In the

commercial sector cost considerations tend to force the use of off-the-shelf commercial instrumentation for automatic testing purposes.

This practice is expected to be positively affected by the recent adoption of standards

(IEEE 488 and American National Standards

Institute CAMAC standards) whose conventions

for interfacing instruments and other devices to computers are beginning to be adopted in the instrumentation industry.

The improvements of semiconductor technology have had a significant impact on the

instrumentation industry in a variety of

ways as shown above. Perhaps the greatest

impact is the growing use of microprocessors

within instruments. Not only are these

small computers used to perform calculations

on raw data to enable the display of useful

information (such as temperature instead of

voltage for thermocouple applications), but

in a more basic function, that of controlling

the internal operation of the instrument.

Contemporary voltmeters and other electronic

measurement instruments function using dedicated logic circuits to control timing,

range changing, and the conversion of analog

signals to digital information. Changes in the mode of operation can only be made by

circuit modification. Use of microprocessors will permit modification of the instrument's operation through software, resulting in more flexibility of design and lower

production costs. Most instrumentation companies are actively engaged in the application of microprocessors to their product

line. At this point in time, announcements

have been made of new voltmeters, counters,

oscilloscopes, and multi-channel scanning

systems, all based on microprocessors.

At the present time, artifact standards,

apparatus and standard measurement techniques to support dynamic measurements of

electrical quantities simply are non-existent. Details of state-of-the-art dynamic

measurement techniques are the proprietary

secrets of the manufacturers of instruments

and signal-conversion devices. As a result,

manufacturers and users of automatic test equipment and systems attempt to ensure the

quality of performance of such systems

through the manual calibration of system

components or the measurement of known static quantities by the system. Neither process gives valid assurance of adequacy of

performance. Such assurance is virtually

impossible without appropriate standards.

The lack of standards also affects the

development of new systems. The designer

may now use published equipment specifications (which in many cases are not uniform

in meaning) to achieve an approximation of

his design goal and then use measurement

results to "fine tune" the system because

standards are not available as design tools.

He may alternatively over-specify in speed

and accuracy to ensure good results. Un- needed accuracy and speed are costly and may

lull the designer to believe he has no analog measurement problems when in fact one may be created by external considerations, such as the impedance characteristics

of the interface between the system and the

unit under test. That the development and establishment

of standards for dynamic measurements of

electrical quantities are a necessity to ease future growth of technology was one of

the conclusions reached during a workshop

held at NBS in September 1974. The workshop,

which had 24 participants, representative of

bpth manufacturers and users, was convened

to identify critical measurement needs and

discuss standards for modern electronic instrumentation. A whole host of standards

and measurement techniques whose need re- sults from the semiconductor and computer

revolutions were identified [13]. They are

best summarized in the abstract of the workshop report, NBS Technical Note 865, given

65
in Appendix C of this report.

The need for a static measurement capability will not diminish in the future. This

is true as static standards must form the basis for uniformity and stability of performance with time for all dynamic instrumentation. The present electrical standards which

provide the basis for accuracy for static

measurements are of materials and combinations of materials inherently more stable

than are semiconductor devices.

In addition, automated equipment is not yet capable of trouble-shooting and repairing

electronics equipment. Although automatic

fault diagnosis for both digital and analog

circuitry is a rapidly expanding area, the

techniques and philosophy are as yet primitive. The construction of present day equipment is such that machine repair would be

prohibitively costly. General purpose maintenance instrumentation must be calibrated

and repaired itself.

Consumer electronics is another rapidly

growing field which will have an effect on

the measurement system. At present, the

main function of the measurement system in

the consumer's market is to enable the maintenance and repair of equipment. However,

the use of electronics for both safety and

energy conservation purposes is growing rapidly in the automobile industry. The potential for disaster due to failure of electronic ignition or braking systems will lead

to more measurements for reliability or quality control either voluntarily, as the

result of government regulation, or as the

economic consequence of successful litigation by crash victims or their survivors.

Another area of consumer electronics

which will increasingly impact the system

is that of medical instrumentation, both by

increasing the amount of support required

and by generating new measurement problems.

Not only is this use of instrumentation in

hospitals and medical research facilities

on the increase, but the summer of 1976 will

mark the beginning of regulation in this

field by the Food and Drug Administration.

Manufacturers of analytical instrumentation

are aware of their liabilities resulting

both from government action and from malpractice suits. A major problem at the present time is the electrical environment in which instrumentation must function. A means of quantifying the "purity" of the

60-Hz power line is needed, both to optimize

the design of instrument power supplies and

to enable hospitals to have proper electrical configurations.

The most important needs for the future

generated by the electronics portion of the

National Electrical Measurement System may

thus be summarized as:

(a) the need for standards for the dynamic measurement of electrical quantities

such as voltage, impedance, and power;

(b) the measurement methodology which

must be used with the above standards;

(c) the education of systems designers

in the necessity for and the use of the above,

before maintenance and failure costs escalate

to render system use impractical;

(d) the establishment of methods and ap- paratus to test computer-driven systems as systems, under conditions of actual use;

(e) the development of a theory to minimize the number of points requiring testing

in a complex system;

(f) a standard and measurement method by

which to assess the quality of ac power

available from the mains;

(g) the characterization of long-term

reliability and stability of electronic

components on the basis of short-term mea- surements (perhaps of parameters such as noise)

;

(h) standards for phase measurements and

for impedance measurements in the frequency

range from 20 kHz to 1 MHz;

(i) the development of improved switches

and relays;

(j) better mechanisms for the communication of technical and management information throughout the system;

(k) a mechanism to ensure that regulatory and other government agency requirements o n the system are technically, economically, and logistically relaistic; and

(1) standardization in the computer soft- ware used for systems implementation of

measurements

.

The electric power industry. This subset

of the National Electrical Measurement Sys- tem requires a considerable measurement

capability, both to cdrry out its day-to-day

operations, and in support of research, the

urgency for which has been created by the

energy crisis. Support of operations re- quires a capability to perform measurements

of voltage, current, and impedance at voltages as high as 765,000 volts, a means of

measuring energy accurately at 60 Hz, mea- surements of transient (lightning impulse,

switching surges) voltages at levels in ex- cess of 2 MV, and the characterization of

dielectrics.

Line-to-line voltages for bulk power

transmission in use or under consideration

by the industry are generally grouped in one of three categories. The hiqh voltage

(HV) group extends from 69 kV (69,000 volts)

to 230 kV. Overhead lines are used for both

short and long distance movement of power

while underground cables are restricted to short distances. A considerable amount of

66
very long distance power transfer (e.g.,

between power companies) takes place on overhead lines at extra high voltages (EHV), some at 345 kV, but most at 500 kV. EHV lines at 765 kV are operational but under

somewhat limited use. One option for power

transfer in the future is by UHV (ultra-high

voltage) lines. Voltages ranging from 1.1 to 1.5 million volts are being considered.

For distribution of electric pov.'er, moderately high voltages between 200 and 45,000

vol ts are employed.

Measurements of voltage and current at these transmission and distribution voltage

levels are performed by means of voltage and

current transformers which reduce the magnitude of these parameters to a level where

they may be handled using conventional low

voltage meters. Both at HV and EHV levels,

inductively coupled transformers are used,

but at EHV there is a trend to use capacitive voltage transformers. Measurement

capability for the calibration of transformers for the HV range is generally adequate

throughout the industry. For the EHV range

the capability of calibrating current and

voltage transformers is generally limited

to the manufacturers of high voltage equipment. The capability of calibrating capacitive transformers (coupling capacitor

voltage transformers — CCVT's) in the place

of installation has recently been developed

at NBS with the support of the Electric

Power Research Institute. The use of high

voltage reactors and capacitors to reduce

circulating quadrature currents in trans- mission systems, and thus, reactive power,

leads to a need for the accurate measurement

of impedance and phase angle for this type

of device. This capability is generally

available only with the manufacturers.

Energy measurements are widely made by

the electric power industry for billing

purposes. This is a very well-established

area; it relates ultimately to the accuracy

of simple, residenti al -type watthour meters

and to more complicated industrial, commercial, and inter-utility metering systems.

There are over 80 million watthour meters

in the ll 4 S., practically all of them rotating, induction-type devices. State utility

commissions legally require that such meters

do not exhibit a certain maximum error, usu- ally 2". The systems by which these meters

are calibrated are detailed in Sec. 4.2.1.

That this portion of the Measurement System

is under adequate control has been demonstrated by means of a round-robin experiment performed jointly by NBS and the

Edison Electric Institute [22].

Testing of the integrity of insulation

of equipment is an important part of the

quality control process of the electric

voltage equipment is subjected to very high

steady state and transient overvoltages

which must be measured with accuracies in the 3 to 5°' range. Such accuracies are difficult to achieve as present instrumentation capable of handling the high voltages required has a poor pulse response

characteristic. A commonly used tran- sient is the standard lightning impulse

(1.2 microsecond front, 50 microsecond tail)

which, for existing transmission voltages,

can be in excess of two million volts.

Another specialized test for the quality

of insulation is that for the level of partial (corona) discharges. The problems here

are the complexity of instrumentation needed and the necessity to isolate the measured

phenomenon from interference. The basic ac- curacy is not a particular problem. Problems

do arise, nonetheless, due to the sophistication of the measurement operation and the

interpretation of data.

The following is a list of representative

measurement problems presently being identified in day-to-day power industry operations:

1. Stability of EHV capacitive transformers is unknown. Hence, their usefulness in monitoring extra-high voltage is limited.

2. Means for calibrating impulse voltage

measuring systems are insufficient. Traceability is usually only through the manufacturer of the equipment. Periodic rechecking

is not widely practiced.

3. There is a lack of convenient and re- liable instrumentation to measure the power

of loads with very low power factors.

4. Measurement and test procedures to predict the life of insulation systems are

i nadequa te.

5. Interpretation of data from corona and

partial discharge measurements requires additional research.

6. There are problems associated with the

very limited market of high voltage instruments. Often they are one of a kind; there

is little opportunity to obtain enough experience to optimize the design.

7. There is need for more in-service testing since many equipment types cannot be re- moved from use for test purposes. In- service measurement methods and instruments

are needed.

8. The industry could benefit from more training of personnel in measurement and

test techniques.

The above problems must be resolved by the

electric power sector in order to identify

components of optimum systems for the future.

The NBS role in solving these is to ensure

that the required measurement capability ex- ists and is available where these problems

are addressed.

67
The energy crisis, the rapid depletion of

petrochemical fuel reserves, and the future

promise of thermonuclear energy have spurred

a diversity of energy-related research efforts [19,23-26]. The effectiveness of many

of these efforts depends on the availability

of appropriate electrical measurement capabilities. In some areas, the key measurements are those generally associated with

the electronics community. For example, the

reliability of control and other auxin iary

electronic systems is recognized as a problem area. The operation and security of

future electric power networks are foreseen

to be more heavily computer-based, and contain large amounts of sophisticated instrumentation for monitoring and control of various parameters [26].

Work is going on in energy storage and

intermittent generation facilities to enable

the main power system generators to handle

only the base loads. Batteries are under

development for storage of energy generated

during off-peak hours for eventual peak-time

consumption. Measurements must be made to characterize the discharge of prototype batteries, as well as to determine their behavior under rapidly varying conditions of

use. Other areas dependent on electrical

measurement are those involving magnetohydrodynamics and fuel cells [26,27].

By observing the current developments in electric energy measuring devices and in metering practices, one may expect early introduction of all-electronic metering systems. This meter may be a more sophisticated device combining the functions of watthour, demand, and power factor meters. It may be coupled to telemetry and computer

systems for automatic reading and time-ofthe-day metering. The accuracy requirements may be similar to those of conventional instruments, but the calibration procedures may be different.

Requirements for measurements at high

voltages come largely from R&D efforts to improve the capacity and efficiency of the

nation's system for the transmission of electric power from generating sites to areas of

use and the distribution of that power within

each area. In discussing these measurement

requirements, the development of future

electric power systems must be considered.

The principal driving forces appear to be:

(1) a continuing increase in the consumption

of electric power, requiring expanded transmission facilities; and (2) concern for us- able systems. The right-of-way available for

overhead transmission will be limited, re- quiring an increase in voltage to transmit

more power. However, there are some doubts

whether further increases will be permitted

because of potentially adverse environmental

effects. There will be pressures to put more transmission lines underground.

Some potential systems, for which re- search and development is underway or under

consideration, are outlined below [23-29]:

Transmission Systems

Overhead

UHV 60 Hz up to 1500 kV EHV and UHV dc up to 1200 kV Underground

Higher voltage tape cables

(765 kV ac)

Higher voltage extruded cables

(to 138 kV and 230 kV ac)

DC cables

SF5~insulated systems

Cryogenic systems

Substations (SFg-insulated)

Underground cables (improve reliability

and efficiency, reduce cost)

Transformers (improve efficiency)

Metering (time-of-day metering, remote

reading, al 1 -el ectronic meters).

There will be requirements for many new types of measurement techniques at high

voltages occasioned by the above. A few of

these are the capability of making accurate

dielectric loss measurements under cryogenic

and other adverse conditions, the measurement

of transient voltages at UHV, the perfor- mance of needed calibrations (transformers,

reactors, capacitors) in an atmosphere of

SFs , and accurate measurements of electric

fields in non-uniform regions for environmental safety purposes. In addition, new calibration techniques will probably be needed

for such new types of measurement apparatus

as electronic wattmeters, capacitive voltage

transformers, and the electro-optical devices which may replace conventional current

transformers at EHV and UHV levels.

In conclusion, the power industry subset

of the NEMS generally functions well, although in several instances the industry

could benefit from improved measurements and' tests. The difficulty lies not so much in the accuracy which can be achieved under

laboratory conditions and in the traceability

to basic standards, but in the very hostile

environment in which the measurements have

to be performed. Many tests are difficult

and expensive to stage because of high voltage considerations and the size of the equipment. Often, measurements are made to elucidate physical processes, e.g., those responsible for insulation aging and deterioration.

In such a case, the problem is frequently

associated with tryinq to relate the measurement results with the physical processes, a research problem on what to measure, not necessarily one of accuracy. Most of the

68
above problems are dealt with by the industry itself or appropriate agencies in the

Federal and state governments. NBS 1 res- ponsibility is for the development of mea- surement techniques and standards needed to solve these problems. NBS has neither the

mission or resources to tackle these problems directly itself.

Looking into the future, one sees great

activity in the development of new and improved electric transmission and distribution systems. There is and will be a need

for new measurements to facilitate the development and to enable the operations of new systems

.

4. SURVEY OF NBS SERVICES

4.1 The Past

Calibration services for basic electrical

standards have gradually improved in accu- racy over the past twenty years. During

that period, the concept of measurement as- surance became the basis for new services as described later in this section. In the

high voltage measurements area some reduction in capability within the NBS facility

resulted from the loss of the Van Ness Street

facility. In 1967, work was begun on a new method of dissemination, utilizing the mea- surement transfer approach. Spurred on by

problems in the area of dc voltage measurements in standards and calibration laboratories, the effort culminated with the establishment in 1970 of the Volt Transfer Pro- gram as an alternative to standard cell calibrations at NBS. In this program, use is made of specially-tested transport standards

to monitor the ability of the client laboratory to make measurements of high accuracy

and precision, as well as to transfer the

unit. The Program has proven to be a powerful tool for the diagnosis of measurement

problems as well as a means for disseminating the volt at improved accuracies. Since

then similar programs in capacitance, resistance, energy, and voltage ratio have been

put into operation.

A new area of in situ high voltage mea- surements stermed from the loss of TBS inhouse measurement capability above 100 kV and the development of portable bridges

based on the current comparator principle.

NBS equipment and personnel calibrate transformers, shunts, and capacitors at the actual site and under the conditions of use.

A similar approach is being taken for the

calibration of high current transformers.

Over the years, the Electricity Division

has contributed a wealth of new standards

and equipment to the measurement industry.

A partial list includes the Wenner bridge

and magnetometer, the' Rosa and Thomas-type

standard resistors, the Hermach thermal

transfer ac-dc voltage converter, the

Cutkosky ac thermometer bridge and the

Cutkosky fused silica dielectric capacitor.

All of these have been adopted for commercial manufacture, or provided significant

contribution to other products.

4.2 The Present - Scope of NBS Services

4.2.1 Description of NBS Services

Structure of the Electricity Division. The

Electricity Division's goal is to provide

within the U.S. the central basis of a com69
plete and consistent system of electrical

measurements, standards, and related data

required by industry, commerce, government,

and the scientific community in their efforts

to advance science, technology, and equity

in trade for the public good. To achieve

this goal the Division has set the following

objecti ves

:

(a) to realize, with continuing improvement, the electrical units and to develop

improved means for their maintenance and

dissemination;

(b) to maintain and disseminate the electrical units and to test and calibrate electrical standards, instruments, and systems;

(c) to develop methods and instrumentation for the solution of problems requiring

electrical measurement;

(d) to determine essential data and constants requiring or related to electrical

measurements when not available with sufficient accuracy elsewhere;

(e) to serve actively in organizations

which set standards for electrical measurements and devices and to otherwise disseminate the Division's specialized knowledge;

and

(f) to evaluate the U.S. system of electrical measurements and to coordinate it

with the systems of other countries to achieve international compatibility.

The basic responsibilities of the

Absolute Electrical Measurements Section are

to relate the U.S. legal electrical units,

the ampere, farad, ohm, and volt, to the

absolute or SI (Systeme International)

units; to develop new and improved reference

standards for the SI units; to develop means

for maintaining surveillance of such refer- ance standards; and to develop improved

means for disseminating the electrical units.

Until recent years, the legal electrical

units have been maintained by NBS using artifact (physical) standards, the stability of

which depended upon materials whose proper*-

ties are generally affected by environmental

changes (temperature, physical shock, etc.)

and impurities, either physical or chemical.

For example, the value of a standard resistor, in addition to being sensitive to tem- perature changes, may undergo a long-term

drift as the result of changes in the intermolecular structure of the wire of which it

is made due to a physical shock and conse- quent deformation. This structure would in

time tend to slowly return to that of an annealed state and the wire resistivity

would reflect this slow change by its drift.

Although so-called absolute experiments,

which relate the electrical units to the

mechanical units, are performed from time

to time, they cannot because of their very

nature be performed often enough nor, for

most parameters, accurately enough to be of use in ascertaining the drift rates of the

artifact standards in which are embodied

the legal units.

It is natural, therefore, that a significant portion of the Section's activity is

taken to exploit processes whose propeties

depend upon fundamental physical constants

for use as primary electrical standards. For example, the U.S. legal volt has been defined

since 1972 in terms of e/h, the ratio of the

electronic charge to Planck's constant. This

is done by the use of Josephson junctions,

weakly coupled junctions of superconductors

which act as precise, linear, frequency-tovoltage converters for which e/h is a con- version factor. Section work currently

includes the development of all cryogenic instruments based on the Josephson effect and

dc current transformer techniques to improve

the quality of the volt (reduce random error)

by an order of magnitude to one part in 10^

and the exploitation of the Josephson effect

for commercially-available laboratory voltage standards at the one part per million

level. Because of the nature of the process,

such a device would never need calibration.

Work in this area and on related voltage

measurements has led to a number of "spinoff" advancements in measurement technology.

Examples include the development of high ac- curacy, sub-millivolt potentiometric methods,

new techniques for the dissemination of the

volt (MAP 1 s ) , application of sophisticated

statistical techniques to standard cell intercompari sons and other measurements, development of automation techniques, and the

design of the world's "cleanest" automated

switching systems.

Present activity in the area of absolute

measurements includes the determination of

the farad and ohm from the unit of length via

the calculable capacitor. This device is ca- pable of producing a change in capacitance

very precisely related to the displacement of

its guard electrodes. After a development

period of a decade, a preliminary measurement of the ohm has been made with an ac- curacy of 0.05 parts in a million [3]. Perhaps equally significant are the developments which were required to achieve the ca- pabilities of maintaining the farad at the

commensurate level of stability and of making

the translation between the unit of capacitance at 5 picofarads, as measured using ac techniques, and the unit of dc resistance at one ohm. New capacitance standards based on the use of a deposited, fused silica dielectric were developed. Although they are inherently quite stable (less than 0.1 ppm per

year drift), their large temperature coef70
ficient required development of temperature

control and measurement techniques at the

microkelvin level of precision. These were

later successfully applied in the construction of a new type of standard cell enclosure. More sensitive amplification and

bridge techniques developed for the necessary

scaling from 0.5 to 1000 pF (10 b ohms at 1592 Hz) are now being employed in the development of automated equipment for impedance

intercomparisons

.

Other measurement activities currently un- derway in the Absolute Electrical Measurements Section include measurements of the

proton gyromagnetic ratio, using the lowfield method; the measurement of the Faraday,

using the silver coulometer; the planning of

new experiments of the ampere and y using

the high-field method; and the exploitation

of cryogenic techniques for the measurement

of resistance standards.

The equipment and facilities available

cover the range from cryogenic and microwave

apparatus to precision dc potentiometers and

ac bridges^ The NBS nonmagnetic facility,

under the jurisdiction of the Section, is available for research requiring a wellcharacterized and controllable magnetic en- vironment. A sputtering laboratory is presently being set up as a tool for cryogenic

and solid state thin film research.

The Absolute Electrical Measurements Section provides the technical base for all of

the measurement activities of the Division.

It disseminates the units of capacitance and

voltage to the Electrical Reference Standards

Section which, in turn, passes needed units

to the other two sections. The latter three

provide the majority of direct technical

services to the National Measurement System.

The Electrical Instruments Section is

responsible for the development of standards

and measurement methodology to support modern, high-speed, electronic instrumentation.

Efforts are underway to provide for the determination of the accuracy of modular ana- log-to-digital and digital-to-analog converters under both static and dynamic conditions. The characterization of high-speed

sampling devices will also be investigated.

Time and speed-related properties which re- quire standards for measurement are risetime,

dynamic or instantaneous input impedance,

settling curve characterization, aperature

time (length and start point), etc. It is expected that the techniques discovered will

"spill over" into the lower speed, high ac- curacy instrumentation area. Previous work

done in the Section on measurements of rms

ac voltage using thermal converter techniques

is being applied to the development of improved ac voltage standards, needed for the

support of modern instrumentation.

Experience gained from earlier work on the development of a sampling wattmeter will

be used to solve two critical problems. The

sampling techniques and mathematical theory

with which to compute the phase angle difference between two alternating signals are

known. Their application to the actual mea- surement situation is being made. The feasibility of using sampling techniques for

the measurement of the power of irregular

waveforms is being studied as well.

The High Voltage Measurements Section

provides measurement support for the electric power industry and its suppliers and users, in addition to conducting research

and development in measurement techniques

and instrumentation for use at voltage and

power levels which exceed those normally

encountered in a standards laboratory.

Calibration services are provided for current, voltage, and energy measurements at power line frequencies and voltages, as well

as for high voltage devices, such as trans- formers, dc and ac voltage dividers, and

impulse dividers. Typically, the Section is concerned with measurements at voltages in

the range from several hundred volts to several hundred kilovolts; reactive power

levels are frequently of the order of

several hundred megavolt-amperes.

The scope of the Section activities is illustrated by the following current re- search and calibration projects:

(a) alternating voltage measurements - voltage (potential) transformers and other

types of high voltage dividers; bridges and

comparators for calibrating voltage trans- formers and dividers;

(b) direct voltage measurements - resistive dividers; generating voltmeters; electrostatic voltmeters;

(c) measurement of transient voltages - electro-optic techniques (utilization of Kerr cell); resistive pulse dividers;

(d) impedance measurements at high voltage - high voltage standard capacitors;

standards for power factor correction ca- pacitors; high voltage capacitance bridges;

self-balancing bridges for measuring losses

in shunt reactors; and

(e) measurement of current in high voltage lines - optical telemetry techniques.

The Electrical Reference Standards Section is primarily responsible for the dissemination of the basic U. S. legal electrical units (capacitance, inductance, resistance, and voltage). These units are disseminated through the calibration in NBS laboratories of standard resistors, capac71
itors, inductors, and voltage cells owned by

organizations from all segments of industry

and government. The units are extended

through the calibration of potentiometers,

bridges, volt ratio standards, inductive

voltage dividers, and other ratio devices.

The units of capacitance, resistance, and

voltage are disseminated additionally by means of cooperative Measurement Assurance

Program (MAP) services for those laboratories

requiring the ultimate in accuracy or desiring to characterize their measurement pro- cesses. These programs employ the use of

extremely accurate transportable standards

and carefully designed experiments to determine the uncertainty with which the client

continues the dissemination process.

The Section also engages in applied re- search work to advance the state-of-the-art

of both the maintenance and dissemination

processes. One example of such work is the

development of the Measurement Assurance

Program for capacitance. The measurement

assurance concept leads to determination of

an all-embracing uncertainty statement, the

product of characterizing the capability of

a client's laboratory to make a specific

measurement. This is done by regarding the

total calibration, maintenance, and dissemination process to any point as a system and

statistically sampling the output of that

system on a basis dictated by the measurement needs of the system. Through the use of NBS transport standards, all random and

systematic errors are determined, including

effects of time, transportation, operators,

equipment, grounding, and ambient conditions.

After satisfactory performance has been documented, information from the system is used

as feedback to reassign the values of the

client's standards to agree with the legal

units. The pursuit of a MAP service for ca- pacitance entailed a study of the transport

and temperature characteristics of commercial capacitors at the 1000-pF level and, as

a result, the design, construction, and testing of a new transport standard.

The Division also serves the National

Measurement System through its committee

activities, consulting efforts, outside-sponsored research and development work, and by

publishing the results of its research and

development work, including that done to support NBS measurement services. Table 12 shows the various technical outputs of the

Division and the approximate level of effort

expended in each. It must be noted that,

because of overlap, the percentages add to more than one hundred.

External sources of program guidance direction are the Office of the Director, IBS; Office of Measurement Services, IBS; the

U.S. Army; the U.S. Navy; the Department of

Defense Calibration Coordination Group (CCG);

Department of Energy (DOE); the Electric

Power Research Institute; the National Conference of Standards Laboratories; the

NAS/NAE/NRC Evaluation Panel; and colleagues

in industry. Since we work in close cooperation on a day-to-day basis with counterparts

in the primary laboratories of the three DOD services, we are kept keenly aware of measurement needs in those laboratories. The CCG

funds projects of joint interest to the

Division and the military calibration structure. Recently, relations have been formalized with the National Conference of

Standards Laboratories which represents the

largest part of the Division's constituency.

Thus, the Electricity Division realizes,

maintains, and disseminates the electrical

units and related quantities to the Electrical Measurement System. These parameters

are disseminated through the Measurement

Assurance Programs and through the calibration of electrical standards and instruments.

The quantities disseminated by the Division

are all derived from three basic maintained

units, those of resistance, capacitance, and

voltage, and the unitless electrical quantities of resistance ratio, impedance ratio,

and ac-dc current difference. The units and

related quantities disseminated and the means

of their dissemination are shown in table

13, page 48. The technical infrastructure has two com- ponents. One is created as a result of the

measurement support requirements of the electronics field and the other is the result of

requirements of the electric power industry.

The requirements of the electronics industry

are of higher accuracy and over a more limited range of voltage and current than those

of the power industry. They cover a wider

frequency range, however, going up to 1 MHz

in some cases. Power companies are generally concerned only with dc, 60 Hz, and

pulse measurements, but at voltage and cur- rents as high as 2 million volts and 50,000

amperes, respectively.

Measurements for Electronics.

(a) DC Measurements.

Voltage measurements of signals ranging

from a few millivolts (thermocouple outputs)

to 100 volts (power supplies for cathode ray and other tubes) are required by the electronics field. The basic unit of dc voltage

is disseminated by the Electricity Division

in two ways. The more accurate is the Volt

Transfer Program in which an NBS transport

standard is used to determine the difference

between the client's unit of voltage and the

legal unit. The uncertainty of this process

72
Table 12. Electricity Division technical outputs

Results and Reports from Sponsored Research and Development 20%

1. DoD CCG (71.5k$)

2. DOE (262. 7k$)

3. Health, Education, and Welfare (FDA/BRH) (5k$)

4. Navy (36k$)

5. Electric Power Research Institute (148k$ - over several years)

6. DoD (20k$)

7. DoD (Army (7.5k$)

Committee Activities 5%

1. IEC (International Electrotechnical Commission)

2. ANSI (American National Standards Institute)

3. IEEE (Institute of Electrical and Electronics Engineers)

4. 0IML (International Organization for Legal Metrology)

5. Internal NBS activities

Consulting 78%

1. Army 5. Aerospace Industry

2. Navy 6. General Industry

3. Air Force 7. Electronics Industry

4. NBS 8. Standards Producers

Calibrations and Testing (Estimated 377k$ FY76 Income) 78%

Industry - Aerospace Government - Navy

(80%) Electronics (15%) Army

Instrumentation

General

Power

Air Force

NASA

Interior (Bureau of Re>

Other NBS Heat Division

Physical Chemistry Division

Office of Weights and Measures

Applied Radiation Division

Optical Physics Division

Electronic Technology Division

clamation)

NOAA

FAA

FDA (HEW)

Bonneville Power Administration

Public Health Service (HEW) EPA TVA

Measurement Assurance Programs (Estimated 45k$ FY76 Income) 12%

1 . NASA

2. DOE

3. Aerospace Industries

4. Power Industry

5. DoD

6. Instrumentation Industry

Publication of NBS-Supported Research and Development 40%

1. Metrologia

2. IEEE Transactions

3. IEEE Proceedings

73
Table 13. Dissemination services of the Electricity Division

Quantity Disseminated Means

DC Volt

Resistance

Capaci tance

Inductance

Ratio

AC-DC Difference

Power

Energy

Volt Transfer Program

Standard Cell Calibrations

Solid State Voltage Reference Device

Cal ibrations

Resistance Transfer Service

Standard Resistor Calibrations

Shunt Calibrations

Capacitance Transfer Service

Standard Capacitor Calibrations

Calibration of Power Factor Capacitors

Standard Inductor Calibrations

Inductive Voltage Divider Calibrations

Current & Voltage Transformer Calibrations

HV Transformer Calibrations

Pulse Divider Calibrations

Volt Ratio MAP Volt Box Calibrations

Resistive Apparatus Calibrations

Resistive Voltage Divider (HV) Calibrations

Thermal Voltage & Current Converter

Cal ibrations

Wattmeter Calibrations

Watthour Meter Calibrations

is about 0.42 ppm. The other way is through

the calibration of saturated standard cells

submitted by the client to the Division for

test. The uncertainty of the test is 0.4

ppm, but transportation effects can raise

that to 1.0 ppm in any context of use. Resistance measurements are used for a large variety of purposes in the electronics

industry. The most stringent requirements

come from the instrumentation industry where

precision resistance components are used in

ratio applications such as ladders (resistive scaling networks) for analog-to-digital

converters and in operational amplifier circuits to provide accurate gain factors. The

unit of resistance is disseminated in two ways from the Division. The Resistance Pro- gram service in which, like the Volt Transfer Program, a set of transport standards is used to determine the difference between the

unit as disseminated by the client and the

legal unit. The Division also provides for

the calibration of standard resistors of decade values between 10~4 and 10'^ ohms. The

uncertainty associated with the Resistance

Transfer Service depends upon the nominal

value of resistors used and the capability

of the client. Transfers at the one- ohm

level have been made with a total uncertainty

of 0.2 ppm. The uncertainty of a regular

calibration is 0.08 ppm at that level, but

transportation effects and errors incurred

in the client's laboratory are not included.

Resistance ratio measurements are used

primarily as tools to scale or subdivide the

units of voltage and resistance. They are

used in the calibration of test equipment

such as digital voltmeter calibrators and

digital multimeters and in the calibration

of other resistive apparatus such as resistance bridges, potentiometers, volt boxes,

and ratio sets. The types of devices used for such mea- surements are potentiometers, volt boxes,

voltage dividers, universal ratio sets, and

Kel vin-Varley dividers. Of these, the last

two are the least specialized. They may be

used to calibrate any of the others. They

can also serve as instruments for two and

four-termi nal resistance measurements and

voltage measurements.

Resistance ratios have the useful quality

that they can be established to very high

accuracies without the use of standards.

Consequently, measurements of ratio quantities are not legally required to be traceable to national standards. However, the

Division calibrates ratio measuring devices

since the facilities for doing so, with very

high accuracies and with a minimum of diffi74
culty, are available. Calibration uncertainties for these devices are generally related

to the resolution of the device (except for

volt boxes) and range from 0.02 ppm to several hundred ppm. The Division also offers the Volt Ratio

Measurement Assurance Program. It is similar in concept to both the Volt Transfer

Program and the Resistance Transfer Service

except that, if measurement difficulties are encountered, a transportable measuring instrument can be sent to the client to be

used as a trouble-shooting device. The un- certainty, due to transportation and measurement, of such a transfer is 1.5 ppm and it

embraces uncertainties due to transportation

and measurement errors at the client's laboratory, as well as at NBS.

(b) AC Measurements.

The unit of capacitance is disseminated

mainly through the calibration of standard

capacitors with values of 10, 100 and 1000

pF at 50, 60, 400 and 1000 Hz. The uncer- tainties of these measurements are 0.5, 0.6

and 5.0 ppm, respectively. The large in- crease in uncertainty for the 1000-pF capacitor is due to the different construction of

the standards used at that level. A typical

standards laboratory should be able to use these standards to make measurements of ca- pacitance in that range of magnitude with no large increase in uncertainty. Such a laboratory would usually have a group of 1000-pF

capacitors as a standard and with normal

equipment should be able to make measurements with a total uncertainty of less than

ten parts per million. This has been borne

out experimentally in tests with military

laboratories.

Inductance measurements are primarily

used industrially for testing of component

chokes and for quality checking of magnetic

recording heads during production. Accuracies required and available in this area are

low. Bridges used to make inductance measurements on the production line are accurate to 1% at best, and the capability of

the Electricity Division to report the values of standard inductors with an accuracy

of 200 ppm is apparently adequate for cur- rent and known future needs.

AC voltage measurements are made primarily with digital voltmeters at the applications level. Accuracies of such meters can

be as high as 0.05% over a range of voltages

from 0 to 1000 volts. They are calibrated

using commercially available calibrators

which are accurate to about 0.02%. These

devices are calibrated by using an ac-dc

thermal voltage converter to permit comparison between their outputs and those of very

well-calibrated dc sources, known as calibrators. The dc calibrators are then verified using standard cells and resistive

ratio devices. The ac-dc thermal voltage

converter is the critical element in this

chain since it permits the determination of rms values of ac voltage using dc voltage

standards. These devices are calibrated by

the Electricity Division with ranging be- tween 5 and 200 ppm accuracies.

AC impedance ratio measurements are primarily used as calibration tools. The most

commonly used device in the electronics field

is the inductive voltage divider. This device is used to subdivide input voltages with

amplitudes as great as 100 volts with a mini- mum adjustment of 10"' of the input voltage.

Such devices are calibrated by the Division

with accuracies of 0.5 ppm of input voltage

(in phase) and 5 ppm of input voltage (quadrature) .

Measurements for the Power Industry. The

power industry used electrical measurements

in the control and monitoring of power. The

main quantities measured are energy, voltage,

voltage and current ratio, and capacitance.

Energy measurements are used to meter en- ergy distributed within the power system for

billing purposes. As stated previously, the

power companies are required by the various

state regulatory power commissions to supply

users with watthour meters accurate to 2%

with a 98% probability. Meter accuracies

are checked by sampling. Higher accuracy, in such measurements can result in smaller sam- ple sizes, leading to considerable savings.

Typically, power companies prefer to make

such measurements to 0.5% to minimize the

probability of fiscal loss by carrying out

tests at higher levels of confidence. A sam- pling of several utility companies revealed

that new meters were well within ±0.5%; old

meters, retested in sampling operations, were within 1%. Meter manufacturers must

accordingly use strict measurement tolerances. There are two routes whereby a utility

can obtain electric energy standards from

NBS. The most direct one is via a client's

rotating watthour meter standard which is sent to NBS for calibration. A variation of

this route is the Measurement Assurance Pro- gram (MAP) whereby an NBS-owned meter is shipped to the client, who calibrates it as an unknown item in his test circuit. This

procedure is preferable since it permits

testing of the client's entire system.

The second, or the dc calibration route,

consists of the client obtaining from NBS

the standards of resistance, voltage, (both

dc) and frequency. In addition, the ac-dc

75
difference of the client's special wattmeter

is also determined by NBS. The standards of

resistance and voltage provide the client

with the standard for the dc power. Utilizing the ac-dc difference of his wattmeter,

he can produce a known ac power. This is integrated using standard time (obtained from

standard frequency signals) to produce the

ac energy. See figure 3. In the past, the dc standards were considered more stable and more easily transferable

than the rotating watthour meter. Establishing an ac energy standard in this manner re- quires more work on the client's part. Until recently, the rotating standard was con- sidered as too fragile a device for transport,

unless hand-carried. During the past four

years, NBS has successfully shipped rotating

standards within the Measurement Assurance

Program without adverse effects. It should

be added that while most of the largest utilities get their standards directly from NBS,

there are utility companies, particularly the

small ones which rely on the standards of

other, uti 1 i ty companies.

High voltage measurements are used for the

monitoring of energy flow in high voltage

lines. This is done by using transformers

to reduce voltages and currents in transmission lines to a level at which they may be

measured by low voltage instrumentation such

as watthour meters.

The accuracy requirements for transformers

in metering service are specified in American

National Standards Institute publications.

The Standard CI 2-1 975, Code for Electricity

Metering, states that corrections for transformer errors do not have to be applied if

the transformers are of the 0.3% accuracy

class (correction less than 0.3%). Another

standard, C57. 13-1968, Requirements for In- strument Transformers, requires that the

equipment for calibration of metering transformers be accurate to 0.1%. No standards

exist for the higher echelon devices - the

reference transformers. The NBS-reported

accuracy of 0.03% for transformers above 100

kV has not resulted in any complaints; nevertheless, a desirability for 0.01% or even 0.002% accuracy has been mentioned by various individuals in the industry.

High voltage impedance measurements are

used to measure the capacitance and loss an- gle of power factor correction capacitors,

and the impedance and phase angle of shunt

reactors. These devices are used to reduce

the reactive power in a transmission system.

Failure to do so would result in power circulating uselessly through the system. Such

devices are calibrated by the manufacturer

using as a standard a gas dielectric capacitor. The manufacturer's measurement system

may be checked out by an on-site calibration

of his standard and bridge using an NBS bridge

and standard. Calibrations of capacitors

rated up to 1000 yF and 100 kVA ratings can

be done at NBS. Appendix B contains a list of primary and

working standards used at NBS in support of

the measurement system.

4.2.2 Users of NBS Services

The users of the services made available

by the Electricity Division fall into the same general categories as do the major com- ponents serviced by the Measurement System.

The major categories are given below.

Table 14. Users of NBS services

Instrumentation Industry

Department of Defense

Aerospace Industries

Communications Systems Manufacturers

Electric Power Industry

Electrical Component Manufacturers

General Manufacturing Industries

Independent Calibration and

Testing Laboratories

Civilian Government Agencies

Universities

User categories are listed roughly in order of their dependence upon the services

offered. As has been stated previously, the

services provide the basis for accuracy of

all measurements which are in turn used to assure the quality, volume, and advancing

technology of produced goods.

4.2.3 Alternate Sources

There are no alternate sources for the

most important service that the Electricity

Division renders the Electrical Measurement

System. The Division provides the research

and development effort necessary to enable

the system to fulfill its needs in a timely

fashion. Other organizations are wellequipped, have competent staffs, and are

able to supply calibrations of standards at accuracies approaching those available at NBS for certain parameters. But none are motivated to make the magnitude of investment required for, nor have the responsibility for providing for, the research and

development in basic metrology needed to support the entire system. No one organization matches the breadth of the NBS capability and none match its capacity for fu- ture improvements or for responding to changing needs. For the measurement system

76
to function in an optimum way, services and

research must go hand in hand.

Commerce and industry today, even more

than at the turn of the century, require

traceability of measurements to a single

source as a necessity for accuracy, without

which interchangeabi li ty of parts, ready re- pair, compatibility of electronic devices,

and ease of design would all be impossible.

In addition, because of its role as the premier source of standards for electro-technology and the fact that it is funded publicly

and represents no special interest, the Electricity Division is able to act as a neutral

third party - an arbiter in areas of technical dispute involving measurements.

Scenarios have been proposed to consider

what would be the results of turning the

bulk of the metrology dissemination and calibration work over to state or private calibration laboratories. To do so would result

in isolation of the Electricity Division from

the bulk of its user community and hence ren- der it less, rather than more, responsive to the needs of the system. The difficulty of

moving technology out to the system would

increase due to this isolation.

4.2.4 Funding Sources for NBS Services

The development of new services and the

improvement of existing calibration services

are supported by congressionally-appropriated

funds (NBS budget), augmented by funding from

private and government agencies outside NBS

on a contractural basis. Typically, the Division has about 20% of its budget provided

in this manner. Most of this comes from the

Department of Defense, to augment work in

support of its measurement systems; and from

EPRI, the Electric Power Research Institute,

and the Department of Energy, for energy-related work. In addition, federal law requires the complete recovery of the costs of

providing calibration, measurement transfer,

and other routine services from the users of

such services. Revenue from this source con- stitutes approximately 14% of the Division's

budget.

4.2.5 Mechanisms for Supplying Services

Dissemination services take place through

three mechanisms: the calibration of client

standards at NBS; Measurement Assurance Pro- gram services, which involve the use of NBS

transport standards to sample the measurement process output at a given place; and in

situ calibrations of such devices as high

voltage transformers and capacitors by NBS personnel using NBS apparatus and standards.

Dissemination of metrology information is provided for by sponsorship of Division sem- inars and workshops, publications in the

scientific literature, participation in standards committee activities by staff members,

publication of tutorial articles such as Technical Notes, and by individual personal

interaction with members of the metrology

fraternity at large.

4.3 Impact of NBS Services

4.3.1 Economic Impact of Major User Classes

Although no accurate quantitative assess- ment of leverage is possible, an appreciation

of the potential impact of NBS services may

be gained by a consideration of the gross annual sales figures of those segments of industry it supports as shown in table 15. The Defense Department, of course, is vital to the well-being of our nation. It depends on measurements to ensure weapons

system reliability, and indirectly through

the aerospace and electronics industries,

Table 15. Electricity Division user classes

Number of Value of Shipments Number of

SIC Code People (billions of $) Companies

CY 75

Electronic Systems 3662 330,000 9.6 86

Components 367 412,000 10.6 1420

Computers 3573 175,000 9.7 610

Engineering & Scientific

Instrumentation 3811 43,000 1.4 721 Measuring & Control

Instrumentation 3821 72,000 2.1 817 Test Equipment 3825 70,000 2.2 622

Power Equipment 351 44,000 2.4 74 Transfo rmers 34433 18,000 .9 40

Electric Power 491 317,000 47.9 225

Aerospace 372,376 615,000 26.2 1225

77
superiority through technical sophistication.

All of these industries, with the exception

of the electric power industry, produce high

technology products and contribute heavily

to our high standard of living. The power

industry applies research and uses high technology to provide us with the means to utilize energy conveniently and is responsible

in large measure for the industrial base of

the country.

4.3.2 Technological Impact of Services

Calibration and MAP services of the Electricity Division provide both the electronic

and electrical power industry segments of

the measurement system with the stable, basic electrical units upon which the accuracy

of measurements throughout the system depends. This support is vital to the development of new, sophisticated instrumentation

and to the maintenance and continuing effective use of the very large existing national

inventory of test equipment and measurement

instrumentation. This inventory of electrical measurement apparatus is the only

viable means of providing quality control

for manufacturing processes throughout industry, ensuring operational readiness of our sophisticated weapons systems, enabling

the high degree of reliability of electronics

and equipment vital to our manned space operations, making possible the reliable and

smooth functioning of the nation's communications and transportation networks, and

providing for the efficient and effective

distribution of electric power to industrial

and domestic consumers throughout the country. The importance of instrumentation and

the measurements it makes is borne out by two factors. The expected increase in test and measuring equipment for electrical quantities alone is 11% between 1975 and 1976

and that growth rate is expected to continue

through 1985, by which time $6.5 billion in instruments will be produced annually [11],

This growth rate, especially in times of

economic hardship, is an indication of the

need for measurements. Secondly, to this

must be coupled the fact that the measurement capability per unit cost of instrumentation is rising dramatically due to the

semiconductor revolution and the inclusion

of microprocessors in individual instruments

[30]. Standards maintained by the Electricity Division and used to disseminate the

electrical units allow measurements made by

this instrumentation to be consistent

throughout the country and stable over time.

NBS services, such as consultative and

committee activities and publication of me- trology research and development work, are aimed at promulgating measurement technology

so that measurements may be made more efficiently, effectively, and accurately throughout the system. NBS representation is most

widespread on those technical committees of

the Institute of Electrical and Electronics

Engineers, the American National Standards

Institute, and the International Electrotechnical Commission which deal with electrical measurements. Literally hundreds of

inquiries per year about various electrical

measurements problems are handled by staff

members

.

Publications concerning measurement techniques have a heavy impact on the measurement community. Over 3000 copies of NBS

Technical Note 430, Designs for Surveillance

of the Volt Maintained by a Small Group of

Saturated Standard Cells by Eicke and

Cameron have been distributed. The measurement techniques set forth in this document

were at the time of publication virtually

unknown in the electrical measurements field

outside of NBS and have since been adopted

by a significant number of metrology laboratories. This has resulted in general improvement in the quality of voltage measurements in standards laboratories as well as permitted the identification of potentially

serious measurement problems.

Some of the areas of technology and industry heavily supported in both development

and operational phases by the measurement

system are semiconductors, discrete electronic components, instrumentation (a user

as well as an impl ementor) ; communication;

navigation; chemistry; metals; and computers.

Electrical measuring instruments provide the

basic tools for physical, biomedical and

chemical research as well.

4.3.3 Pay-off from Changes in NBS Services

Two notable changes have occurred in re- cent years in services offered by the Electricity Division. The first is the increasing use of Measurement Assurance Program

services, starting with the Volt Transfer

Program, for the dissemination of the electrical units. In the case of all of these

services there have been two immediate technical effects: the units have been transferred with higher accuracy; and invalid

measurement techniques at users' locations

have been identified and their use curtailed

due to NBS analysis of data.

In a number of cases, there has been a third payoff. The techniques used in the

process have been successfully applied to the users' individual measurement system.

The most definitive economic benefit has

been the extension of the interval between

calibrations for the clientele using these

services. This is made possible by the

78
improved accuracy of the transfer and the

methodology by which the data are reduced,

in comparison with usual methods used to ensure measurement traceabil ity. The

greatest measurable economic impact of this

type of service will most likely result from

the application of these specific techniques

to effect a reduction in cost of supplying

measurement support to production lines and

throughout quality control systems.

For a group of five California companies,

the availability of MAP services has meant a completely new method of establishing traceability. In the past, each company sent standards of voltage and resistance annually

to NBS for intercomparison. Now, each year

only one of the companies becomes involved

in an NBS voltage or resistance transfer.

During the same period of time, the group

undertakes a five-way intercomparison of the

appropriate units for each company, using

measurement transfer techniques. They have

reduced the cost of traceabil ity significantly in this fashion and each now has con- fidence in the standards of the others

should a failure occur in his own £31],

The second major change in the Division's

services occurred as a result of closing the

old NBS high voltage (HV) laboratory in Washington, DC. In 1970, a decision was made and implemented to move, the low and intermediate voltage support facilities and

offices of the High Voltage Measurements

Section to the Gaithersburg site. The HV laboratory was intended to be retained as a field station and used whenever the need arose for voltage levels above 50 kV. However, in 1971, the General Services

Administration, the owner of the laboratory

and site, asked NBS on short notice to va- cate the facilities in order to make the

land available for the Washington Technical

Institute. NBS abandoned the laboratory and

transferred the equipment for a nominal sum

to another government agency. Thus, NBS

gave up, and the Division lost, a facility

whose replacement cost today would range

from $5 to $10 million.

The basic reasons for this closing were

complex: low calibration workload, a lack of

funding of high voltage research by other

agencies; its vulnerable position as an isolated facility away from the main NBS campus; and NBS program priority decisions.

To retain some capability for calibrations and research in the high voltage mea- surements area, especially for measurements

required in revenue metering, a new mode of

operation was begun. Instead of doing all

HV work in-house at NBS, measurement services

above 100 kV are offered as on-site calibrations. Division staff members go either to customers' high voltage laboratories or to

one which belongs to an acceptable third

party. The NBS instruments and standards

used are designed with portability in mind.

For calibrations and research below 100 kV,

and intermediate voltage in-house facility

has been established at the NBS Gaithersburg

(MD) location.

This change in NBS facilities extensively

affected the electric power industry. While

retaining an acceptable calibration capability, it reduced the Division's capability

and flexibility to contribute to the research

requirements of electric power transmission

systems. The level of power industry support in the sixties, when the future of the

NBS high voltage laboratory was debated, may

not have indicated the need for it. However, the situation changed significantly in the past few years.

The principal reason is the energy crunch

and the associated emphasis on energy-related R&D. Electric power R&D in industry has

been focused by the formation of the Electric Power Research Institute (EPRI), which

is the R&D arm of the power industry; and

the Department of Energy, which controls the

public funds related to energy research.

The combined transmission and distribution (T&D) budget for these organizations is about $40 million annually. The size of the

measurements component is difficult to assess exactly since it is diffused throughout

the entire program, but a safe estimate,

based on the problems of which NBS is aware, would be at least 1% of this or $400k annually.

In the past two years, power systems re- lated OA work in the Electricity Division

has increased from a negligible amount to about $250k annually. Further increases

still could be made if facilities and man- power were available.

In summary, the potential need for mea- surements R&D related to electric power systems is at least $400k annually. Even if

other constraints were removed, we could not undertake such a program fully without having access to our own or a closely located

HV laboratory. It also must be added that

T&D research (including measurements R&D)

relates to a projected capital investment of

$150 to $200 billion over the next 15 years.

79
4.4 Evaluation of NBS Program

The Electricity Division program is generally consistent with the needs of the electrical measurement system but inadequate in

several areas, such as the dynamic electrical standards area, due for the most part to limited resources, both of personnel and

funding. The most powerful tool for determining the needs of the measurement system

for the Division's services and output is the expertise of the technical staff of the

Division. By keeping current with trends

and advances in instrumentation and electronics, staff members identify potential

areas where NBS measurement support will be

needed as well as up-to-date tools to help

supply that support.

Other agency sponsors constitute a powerful means of identifying needs. In the electrical power area, responsibility for the

development of nearly all future technology

is in the hands of the Division's two major

sponsors, the Department of Energy (government) and the Electric Power Research Institute (private sector). Widespread Division

contacts within these two organizations en- sure early recognition of needs within the

power industry.

State-of-the-art advancements in electronics, systems, and instrumentation generally

result from fulfillment of the new weapons

requirements of the Department of Defense.

The Division has a very close coupling with

the Defense Department through its Calibration Coordination Group. This DoD group,

which over the years has sponsored a large

amount of metrology development work in the

Division, apprises NBS staff members of future measurement requirements in the military system.

The National Conference of Standards Laboratories is a management-oriented organization comprised of companies and government

agencies with metrology interests. This

group is highly representative of the Electricity Division's clientele as there is

greater than a 80% overlap of NCSL member

companies and the Division's calibration ser- vices users. Recently, the Executive Board

of that organization approved a plan to con- stitute an NCSL liaison board to the Electricity Division. This board will serve both

to provide inputs for future planning in the

Division and as a vehicle for the dissemination of technical and management information

from NBS to NCSL member companies.

The National Academy of Sciences/national

Academy of Engineering/National Research Council Evaluation Panel for the Electricity Division reviews the programs in the Division,

generally on an annual basis, and provides

valuable input for the determination of future program directions. The most recent

such panel had eight members representing

the viewpoints of the power industry, the

Federal energy research program, the Department of Defense, the academic community, the

electronics industry, the research community,

and the instrumentation industry. The Pan- el's reports contain comments and recommendations on technical, managerial, and administrative matters as well as suggestions for

new directions. The annual meetings between

the Panel and staff members, although perhaps too brief, are very beneficial for the

exchange of technical information and views

too detailed to be of use in management re- ports . The most recent Panel was in full support

of the Division's ongoing programs, and es- pecially highlighted the need for work to be

done in the areas of measurements in support

of high voltage research and development and

high-speed dynamic standards for modern electronic instrumentation. The Panel identified

communications between the Division and its

constituency as being a major problem and

commented on the process of setting long- term development goals in the rapidly developing area of modern instrumentation.

As needs are identified, program priorities must be established. These priorities

are the result of careful consideration of

the degree to which response to the need

falls within the NBS mission, the magnitude

of the need, the resources (including expertise) available to deal with the problem,

and the likelihood of success in an appropriate time frame.

Major needs of the system for NBS services

include those for:

(a) basic standards to maintain stable na- tional electrical units;

(b) means of disseminating the units;

(c) "how-to-to-i t" metrology information;

(d) dynamic measurement standards for modern instrumentation;

(e) techniques for the measurement of impedance, voltage, current, and dielectric

properties at high voltages; and

(f) the determination of fundamental physical constants, such as the Faraday and 2e/h.

The Electricity Division has on-going programs to respond to each of these areas of

need. Because of the widespread need of

these services and the need for an impartial,

technically competent organization to deliver

them, it is clear that they can be performed

only by a federal government agency. NBS is

the only agency with the expertise and statutory responsibility for providing them.

80
4.5 The Future

Activities in the Electricity Division

are at present undergoing a period of change.

Advantage has been taken of the completion

of a large number of projects in traditional

areas to effect reprogramming to bring effort

to bear on the solution of problems in new measurement areas. The Electricity Division's

capability to provide support for static mea- surements in the general instrumentation are generally more than adequate to meet present

needs. However, similar support is lacking

for the dynamic measurements (i.e., those

made under a time constraint) of electrical

quantities such as are made under computer

control. Hence a considerable effort is being devoted to a program to establish measurement methodology and standards in support of

modern, high-speed instrumentation. Investigations are proceeding to establish standards

and instrumentation to investigate high-speed

analog-to-digital and digital-to-analog con- version modules. A program to establish methodology for the proper support and implementation of electronic test equipment of high

accuracy is underway on a modest scale. The

Division is midway in the development of a system to provide an accurate measurement

base for the semiconductor industry at the

10-volt level

All of this activity is in response to the

needs perceived by Division personnel which

led to a workshop in September of 1974 to identify and quantify Critical Electrical Mea- surement Needs and Standards for Modern Electronic Instrumentation. The Workshop was attended by 24 engineers and scientists representing a variety of industrial viewpoints.

Automatic test equipment users, instrument

manufacturers, systems users and manufacturers, and the semiconductor module industry were represented, mostly at the engineering

manager, or vice-president level. The general conclusion of the Workshop supported the

Electricity Division's perception of needs

in this area. Other hitherto unidentified

needs were made known, most notably in the areas of pulse responses of components, dielectric hysteresis, and the characterization of ac line voltage with respect to transients, dynamic impedance, etc. [see Appendix

C].

In the area of dissemination of information, a program has been started to generate

a series of tutorial publications covering

various aspects of electrical metrology useful to the personnel of a typical standards

or calibration laboratory. An effort is being made to establish an Electricity Division newsletter in one technical journal and

to publish overview articles in other, more

generally based, periodicals, in order to

expose Division programs to the scientific

and engineering community and thus engender

feedback.

In discussing the measurements for electric power systems this report implicitly

and explicitly recognizes two areas: (1) the areas related to the "traditional" quantities

such as voltage, current, impedance and, es- pecially, power and energy; and (2) that area related to research and development of new apparatus and systems, primarily for transmission and distribution. In the latter area are also included those specialized and difficult measurements employed in the design

and quality control of present systems.

The delineation between the two areas is not very rigid and the term "traditional"

must be tempered since it does not imply lack

of need for research and development activity

As the instrumentation and transmission technologies change, NBS must meet the changing

measurement requirements for these basic

electrical quantities. As discussed elsewhere in this report, a very important item

in this first category is the measurement

of electric energy. Although currently this area is well in control, new instrumentation

and metering practices will be introduced

within the next five years which will place

additional requirements on the NBS services.

Electronic watthour meters are appearing in

interutility and large commercial and industrial metering installations. Load management based on variable rates for electric

energy is being considered. A more sophisticated instrument combining the functions

of a watthour meter and a demand meter has

been suggested. As with the conventional induction-type watthour meters, not only will

the accuracy and reliability be of concern,

but also the ease of remote reading and control and interfacing with a central computing

facility. Although its role is not entirely

clear at the moment, NBS will be required to provide rather different kinds of supportive

measurement services in the future in addition to the present routine calibrations and MAP services. New developments in this area

will be closely watched.

In measuring the traditional steady state

quantities, problems exist in calibrations

at higher voltages, where relatively few or- ganizations have a capability of calibrating

voltage transformers. Traceability to the

fundamental NBS-mai ntai ned standards is weak.

The Electricity Division will try to alleviate this situation by promoting field calibration services and through specific projects such as the development of the system

for calibration of coupling capacitor voltage transformers in substations.

With respect to the other area concerned

with R&D and specialized measurements, there

81
are numerous unresolved problems now and cer- tainly more will become evident as the R&D

in new transmission systems progresses. The

weakest area now is concerned with the mea- surement of high voltage transients, especially at extra high voltage (EHV) and ultrahigh voltage (UHV) levels. NBS will continue

to work in impulse measurements with particular emphasis on transferring our capabilities to the power industry. Contingent upon

the availability of resources, work should

be done on devices for measuring voltage,

current and power in high voltage dc trans- mission systems.

The Division will follow the developments

of new transmission systems and provide its

expertise for solving measurements -related

problems. This area is proper for extra-NBS

support. An example is the development of

a superconducting ac transmission system.

High voltage measurements research related

to the insulation of superconducting cables

is already being performed. Superconducting

systems will have very high current capabilities. New methods will be needed to measure these currents.

Another new transmission system under

development uses SFg-insulated conductors.

Nonconventional measuring methods appear to offer cost and accuracy advantages in these

systems. This entire area is ripe for ex- ploration.

If overhead transmission enters the UHV region, electro-optical current transformers

will become very attractive. The NBS will

be in the best position to evaluate their ac- curacy capabilities by virtue of measurement

techniques in the UHV region developed here.

Potentially adverse effects on environment due to high electric fields, audible

noise and electromagnetic interference are

of great concern in connection with new EHV and UHV overhead systems. The Division will

offer measurement assistance to the agencies

primarily concerned with this problem.

Reprogrammi ng efforts to respond to all of these needs made evident a serious weakness in the Division. The Division simply

lacks the proper resources, people, and

funding to mount a comprehensive program in

each area without seriously eroding the supporting basic measurement program - there is concern that a proper job cannot be done

without ignoring clear responsibilities in

the traditional areas.

On the other hand, there would be a large

negative impact if the Division were to fail

to meet the needs being addressed by these

various activities. The state of the electronic marketplace is one of chaos with re- spect to high-speed electronics. Lack of common terminology and universally accepted

test procedures makes procurement by specification a difficult and hazardous procedure.

Since the definitions of temporal parameters

are nebulous, different interpretations are

used by different manufacturers and the buyer

has no means of evaluation except in terms of a specific application. The net result

is that much money is wasted on useless purchases and the over-specification of components to ensure adequate performance. The

presence of an authoritative, technically

competent, neutral "third party" such as NBS,

is urgently needed in the area to establish

commonal i ty.

Why should NBS be involved in any of

this? The Federal government is the only or- ganization that can afford to look past to- morrow to the future in the area of measurement science and standards. Industrial con- cerns must have staying-in-business as a first consideration. That generally means a minimum of long-range (10-20 year) research

programs, and a great deal of motivation for

keeping potential profit-making results of

such research secret. The National Bureau of Standards is the

only government organization with the base,

the expertise, and the legislative charter

to perform these functions. No other government laboratory has appropriate facilities or staff. Research in measurement methodology

would not be the primary concern where other

national needs such as national defense are

the responsibility of the organization. The

country needs one organization in which mea- surement activities and standards are the

primary concern as at NBS.

82
5. SUMMARY AND CONCLUSIONS

The National Measurement System for Electricity has two major components; those of

electronics and the electric power industry.

These are separately identified because of

the distinctly different nature of the r surements required in each. The former is comprised of the electronics and instrunentation industries and their users, while the

power subset of the system is comprised of

the electric utilities, manufacturers of

generating, transmitting and distributing

apparatus and, to some extent, the users of

electric energy.

The present measurement capability of the

electronics subset is more than adequate to support traditional measurements (those mad'

manually or without time constraint) except

in a very few limited cases such as, low 1 .— quency, low amplitude ac voltage measurements

and upper audio and lower radio frequency impedance measurements. However, due to the

impact of advances in semiconductor technology on the instrumentation, computer, and

electronics industries, and the resultant

increases in complexity of components, devices, and systems, a larger and larger percentage of measurements is being made by

electronic instrumentation under computer

control and under conditions in which time is

a constraining factor. This leap forward in

measurement technology has not been matched

by the development of appropriate standards

of measurements. Designers of automatic

test systems and equipment do not have the

benefits of either physical or written standards for dynamic electrical measurements as tools. Common terms, such as aperature and

settling time, used to describe dynamic p*"

erties of measurement equipment, do nc' universally accepted definitions. A physical standards, analogous to standard resistors, capacitors, and cells, for the quantification of dynamic properties do not exist.

Designers and users of modern test equipment

and instrumentation must use costly trial

and error or "overkill" methods for both the

development and maintenance of such equipment. Development of future, more accurate

and faster systems will be hampered by lack

of such standards.

In the electric power industry subset,

which deals basically with energy, voltage,

current, and impedance measurements at 60 Hz, some at very high voltage and impulse measurements at high voltages, the present ca- pabilities are generally adequate to support

today's operational requirements. However,

the electric power industry is beginning an

era of great change. The main forces for

change are the energy crisis and the appreciation of the small size of the remaining

supply of petroleum. Research efforts to improve the capacity and efficiency of elecpower generation, transmission, and distribution have been expanded greatly over the

past few years. New measurement methods

and standards are drastically needed to en- able this research to reach fruition and to support and maintain the new power systems

expected to result. Measurement methods

are needed for such things as calibration of

capacitive coupled voltage transformers

(CCVT's) for use at 500 kV and above; evaluating various types of prototype distributions systems, such as underground dc and gas

insulated, cryogenic, or extruded insulation

systems, and overhead systems ranging as high

as 1.5 million volts; evaluation of environmental and interference effects of ultra-high

voltage overhead lines; and development of

tests to ensure that equipment on such lines

can withstand transient overvoltage and surge

currents caused by lightning and switching.

Energy metering, done largely at the present

time using rotating electrodynamic meters,

will require new measurement support with the

advent of electronic meters which will be

used for demand and time of day metering purposes and may eventually employ telemetry

techniques for instant transmission of con- sumption information to the utilities.

There are a few common problems with widespread effects on the entire system. Among

them are:

(a) a lack of availability of "how-to-doit" measurement information throughout the

system;

(b) a shortage of manpower with specific

metrology training and a resulting widespread

lack of sophistication in measurement technology as practiced;

(c) heavy emphasis on "traceabil ity",

largely fostered by Department of Defense

quality requirements, which causes much attention to be focused on equipment specifications

and calibrations rather than measurement technology. This tends to slow the development

of new and improved methods of providing mea- surement support where it counts, on the production line or at the weapons system; and

(d) a lack of recognition of the formal

metrology structure as an asset rather than

a required artifact in a large percentage of

industry.

Specific recommendations resulting from

this study are:

83
General In support of the power industry

Develop a mechanism to foster and support

the use of Measurement Assurance philosophy

and techniques within industry to provide subsistence to measurements made for quality control, manufacturing, or maintenance purposes.

Develop more formal contact between the

Electricity Division and the instrumentation

industry, perhaps through the aegis of the

Scientific Apparatus Manufacturers Association.

Establish national standards and improved

measurement methodologies for the ac measurement of conductivity in metals in support of

non-destructive testing in industry.

In support of electronics

Investigate new technology for providing

measurement support for automatic test systems, perhaps in conjunction with Defense

Department efforts in this area.

Examine microprocessor technology and the

applications of microprocessors to instrumentation to lay the foundation for support for

future generations of sophisticated "smart" instruments.

Provide improved methods for the measurement of ac voltages at low amplitudes in the

sub-audio frequency range and the measurement

of impedance in the upper audio range for the

support of vibration and communications instrumentation respectively.

Provide for development of measurement

methodologies for the characterization of the

insulating properties of various materials

and systems at high voltages and under various conditions and configurations (such as for superconducting cables).

Contribute to the development of electronic

watthour meters to develop the new standards

and measurement techniques necessary to their

support.

Promote the use of field calibrations and

measurement assurance programs in the area

of high voltage.

Continue research in high voltage impulse measurements and develop mechanisms

to disseminate the resulting technology to industry.

Develop new approaches and devices for the

measurement of voltage, current, and power in

high voltage dc transmission systems.

Investigate methods for measurement of the

extremely high currents which will be carried

by superconducting cables.

Develop methods for the precise measurement of electric fields in regions of extreme

nonlinearity for environmental protection and

safety testing purposes.

84
APPENDIX A

THE MEASUREMENT SYSTEM STUDY

The starting point in efforts to date has

been the group of organizations constituting

our calibration clientele. The general plan

for this study is outlined below:

a. Identify calibration clients and

associated economic data.

b. Classify clients by industry or major

product(s)

c. Select sample of clients to visit to determine measurement needs, technical problem areas, and organizational structure.

d. Identify corporations and industries

who are not clientele but who are engaged

in the same kinds of endeavors.

e. Sample non-client organizations of

maximum economic, social, and political importance to determine the methods employed

to achieve product quality "surance in

those organizations.

f. Garner detailed in cion about

specific needs via intervie. , workshops,

etc.

Using information supplied by the Office

of Measurement Services (NBS), a listing of

all users of our calibration services in FY

72 has been compiled. The clients on this

list were then classified in one of the categories shown subsequently. Each organization was assigned up to three numbers

representing one primary and two secondary

activities. These were based on personal

knowledge and on descriptions found in Standard and Poor's Index. In addition, annual income figures from either the Standard

and Poor's Index or the Dun and Bradstveet

Million Dollar and Middle Market Indicies

were gathered for each customer. This information was added to that supplied on punch

cards by the Office of Measurement Services

and sorted. There were three sorts made,

one by each of the primary and secondary

activities of each company. As a result of

the first sort, 52 companies are unclassified; 17 are in the aerospace industry;

49 manufacture instrumentation; 21 manufacture components; 43 are power industry-related; 7 are universities; 9 are other

Federal government agencies; and 32 are in general manufacturing, for a total of 230

companies. A meeting of the study authors

was held on the receipt of these printouts

and a number of pertinent decisions were

made.

It was decided that a separate substudy

should be made of the power industry in view

of the percentage of the Division's clientele

it represents, the annual income it has, the

role it plays in our society, and the energy

crisis. The chief of the High Voltage Section undertook this substudy because of the

mission of the Section and his personal contacts within the electric power industry.

A minimum sample size for each of several

categories was decided upon, based upon the

perception of the degree of impact NBS has

on each category and knowledge of the uniformity of measurement technology within the

category. The table below gives each of

these categories and the sampling size

chosen.

SAMPLE SIZE FOR CLIENTS

Category Sample Size

201 Precision Apparatus 100%

212 DC-LF Test Equipment 100%

100 Aerospace Industries 25%

209 Systems 10%

600 Other Agencies 100%

300 Component Manufacturers 2 of each subcategory

800 Calibration Laboratories As many as possible

The type of information which would be

sought of each sample was also discussed.

Answers to the following questions would be

sought:

a. What are the corporate quality control

organizational and technical structures?

b. Do you perform calibrations for other

organizations? How Much? For whom?

c. What is the name of the key researcher

in your company (Chief Scientist, perhaps)?

d. What is the annual expenditure on electrical measurements in your company?

How is it spent?

e. What measurement areas are most critical to your final product?

f. What measurement or measurementrelated problems do you have? Can NBS help?

g. What products are most critically

affected by measurements?

h. Would you be aided by improved or new

NBS services? What areas? Why?

i. What are the desired accuracies of

measurements at the end item? Who determines

these? Why are they what they are? What are

the consequences of not supplying the desired

accuracies?

j. Are you a military contractor? Are you subject to any regulatory agency policies?

85
It also seemed desirable to inform our contact, in advance of a visit by Division

personnel, of the capabilities of the NBS and

especially of the Division. Accordingly, an effort has been made to inform him by letter

or telephone of the specific purpose of the

visit so he could be prepared. It would be

difficult, otherwise, to qet past the cor- porate standards laboratory into the ap- plied areas of the system.

In gathering information from Standard

and Poor's Index and from the two publications of Dun and Bradstreet, it became

apparent that NBS has direct contact with

only a small percentage of the electrical

measurement system. It had been originally

decided that the Division must sample that

part of NEMS it does not normally interact

directly with. To do that the services of

Dun and Bradstreet were enlisted. The list

of clients previously mentioned was expanded

to include FY 73 clients and each plant location served. The local office of Dun and

Bradstreet determined the primary Standard

Industrial Classification (SIC) Code for

each company of a large sample. After discarding 17 of the 35 SIC codes thus obtained

due to obvious lack of relevance, a contract

was issued to Dun and Bradstreet to supply

the Division with Dun's Market Identifiers

for each of the companies in their file

having one of the 16 primary SIC codes and more than 20 employees. Information has been

supplied for 12,865 companies in the United

States,

We hope to obtain additional samples by

interrogating people who visit the Division.

In the past four years over 90 plant locations have been visited by Electricity Division personnel. Before this study was formally initiated, the purpose of most

visits was the discussion of measurement

problems. The next pages contain a representative list of locations visited.

Technical forecasting is always involved

in program planning, whether explicitly or implicitly. In the past, technology forecasting for this Division has been done by two groups of experts, one (unorganized)

within the Division and the other outside

of the Division, the NAS/NRC/NAE Advisory

Panel

.

The technology forecasting portion of

this study has been accomplished by doing a formal review of trends and forecasts already published for the electrical field by

trade and industrial associations, professional societies, and other government

agencies.

SAMPLE LIST OF PLACES VISITED

Associated Engineering Company

Matthew, NC 29105

Autonetics

Rockwell International

3370 Miraloma Avenue

Anaheim, CA 92803

Ball Brothers Research Corporation

P. 0. Box 1062

Boulder Industrial Park

Boulder, CO 80302

Baltimore Gas and Electric Company

5 31 E Madison Street

Baltimore, MD 21203

Beckman Instruments, Inc. 2500 Harbor Boulevard

Fullerton, CA 92634

Bendix Environmental Science Division

1400 Taylor Avenue

Baltimore, MD 21204

Boeing Company

Aerospace Division

P. 0. Box 399

Seattle, WA 98124

Brookhaven National Laboratory

Brookhaven, NY

Dickson Electronics Corporation

P. 0. Box 1390 Scottsdale, AZ 85252

Duke Power Company

P.O. Box 2178

Charlotte, NC 28201

Fairchild Semiconductor

4300 Redwood Highway

San Rafael , CA 94902

Ford Motor Company

21500 Oakwood Boulevard

P. 0. Box 1704

Dearborn, MI 48121

General Electric Company

Ordnance Systems

100 Plastics Avenue

Pittsfield, MA 02101

Global Associates

Mississippi Test Facility

Bay St. Louis, MS 39520

Hewlett Packard Company

815 SW 14th Street

Loveland, CO 80537

86
Hewlett Packard Company

Automatic Testing Division

Mountain View, CA

Hewlett Packard Company

Microwave Division

Palo Alto, CA

Honeywell Industries

5558 Port Royal Road

Springfield, VA 22151

Honeywell, Inc. Government & Aeronautical Products Div.

2600 Ridgway Parkway

Minneapolis, MN 55413

Honeywel 1 , Inc. Test Instruments Division

4800 East Dry Creek Road

Denver, CO 80217

Honeywell, Inc. 5303 Shilshole Ave., NW Seattle, WA 98107

Hughes Aircraft Company

Centinela & Teale Streets

Culver City, CA 09230

IBM Systems Manufacturing Division

Monterey & Cottle Roads

San Jose, CA 95193

Jet Propulsion Lab

California Institute of Technology

4800 Oak Grove Drive

Pasadena, CA 91103

John Fluke Manufacturing Co.

P. 0. Box 7428

Seattle, WA 98133

Julie Research Laboratories

211 West 61st Street

New York, NY 10023

Leeds & Northrup Company

Sumneytown Pike

North Wales, PA 19454

Linde Corporation

Tarrytown, NY

Lockheed Missiles & Space Company

P.O. Box 504

Sunnyvale, CA 94088

Martin Marietta Corporation

Denver Division

P. 0. Box 179

Denver, CO 80210

Melpar

Division of LTV El ectrosy sterns 7700 Arlington Boulevard

Falls Church, VA 22046

3M Company

3M Center

Bldg. 21 6-1

St. Paul , MN 55101

Newark Aerospace Guidance & Metrology Center

Newark Air Force Station, OH 43055

Optimation Industries

Northridge, CA

Patuxent Naval Air Station

Patuxent River, MD 20670

Oak Ridge National Laboratory

Oak Ridge, TN

Philadelphia Electric Company

2301 Market Street, N4-1

Philadelphia, PA 19103

Rosemount Engineering Company

12001 West 78th Street

Eden Prairie, MN 55343

Shallcross Manufacturing

Preston Street

Selma, NC 27576

Sperry Rand Corporation

UN I VAC Division

Univac Park

P.O. Box 3525

St. Paul, MN 55101

TRW Systems Group

One Space Park

Redondo Beach, CA 90278

United Air Lines

San Francisco International Airport

San Francisco, CA 94128

Westinghouse Electric Corporation

P. 0. Box 1637

Bal timore-Washington International

Ai rport

Baltimore, MD 21303

Westinghouse Electric Corporation

Astronuclear Laboratory

P. 0. Box 10864

Pittsburgh, PA 15236

Westinghouse Electric Comply

Raleigh, NC

87
CATEGORIES OF CALIBRATION CLIENTELE

100 Aerospace Industry

101 Prime Contractors (Airframe)

102 Subcontractors (Miscellaneous)

103 Subcontractors (Propulsion)

104 Subcontractors (Guidance)

105 Consultants

200 Instrumentation

201 Precision Apparatus Manufacturers

202 DC-LF Test Equipment Manufacturers

203 Analog-to-Digital , Digital-to-Analog

Conversion Equipment

204 Medical Instrumentation

205 Process Control Computer and Data Acquisition Products

206 Nuclear Instrumentation

207 Communications Equipment

208 Recording Equipment

209 Systems

210 Chemical Instruments

211 Pollution Monitoring Equipment

212 Microwave Test Equipment

213 Temperature Measuring Equipment

300 Component Manufacturers

301 Resistors

302 Capacitors

303 Inductors

304 Solid State Reference Devices

305 Integrated Circuits

306 Switchgear

307 Electron Tubes

308 General

309 Batteries

400 Power Industry Related

401 Transmission Monitoring Equipment

402 Generating Equipment Manufacturing

403 Power Measuring Equipment

404 Power Generation

405 Consultants

500 Universities

600 Other Government Agenices

601 Army

602 Navy

603 Air Force

604 NASA

605 Bonneville Power Administration

606 Public Health Service

607 FDA 608 N0AA

609 FAA

610 EPA

611 TVA

612 Bureau of Mines

613 ERDA

700 General Industry

701 Automobile Industry

702 Waste Management

703 Electric Motors

704 Electric Generators

705 Metals Products

706 Nuclear Research

707 General Research

708 Computer Manufacturing

709 General Manufacturing

710 Chemicals

711 Consumer Products

712 Services

713 Sales

800 Calibration Laboratories

88
APPENDIX B

NBS PRIMARY AND WORKING ELECTRICAL STANDARDS

The following identifies the number and type of standards used to maintain the various

electrical units and working standards for the dissemination processes. The term "primary"

as used below refers not only to those standards used to maintain the basic electrical

units, but to those which enable measurements to be made under conditions differing from

those under which the basic units are maintained. Examples of the latter are standard

resistors especially designed for stable characteristics at very high currents. The list

does not contain measurement apparatus (except when such apparatus embodies a unit), or intermediate standards used for scaling, or the check standards used to maintain statistical

control over the various measurement processes.

Sec. 1. RESISTANCE

Primary Standards

5 Thomas (NBS) One-ohm Standard Resistors

Working Standards used at Low Currents (0.01 W or less)

10

10

10

10

1010

1010

10

10

10

1010

10

10

10

3

10*

4 -3

2

-1

0

12

22

34

45

6

ohm

Primary Group)

Resistors (MAP)

io10

10

10

1010

10

10

11

1213

14

15

4 Rei chsanstal t (commercial) Standard Resistors

4 Reichsanstalt (commercial) Standard Resistors

4 Reichsanstalt (commercial) Standard Resistors

4 Reichsanstalt (commercial) Standard Resistors

5 Thomas (NBS) One-ohm Standard Resistors (The

15 Thomas (commercial and NBS) One-ohm Standard

4 Rosa (commercial) Standard Resistors

2 Special, high-stability (commercial) Standard Resistors

2 Special, high-stability (commercial) Standard Resistors

2 Special, high-stability (commercial) Standard Resistors

2 Special, high-stability (commercial) Standard Resistors

4 Special, high-stabil ity (commercial ) Standard Resistors

4 Special, high-stability (commercial) Standard Resistors

4 Rosa (commercial) Standard Resistors

4 Rosa (commercial) Standard Resistors

2 Wirewound (commercial) Standard Resistors

2 Film (commercial) Standard Resistors

2 Series/parallel (NBS) Resistance Boxes

2 Wirewound (commercial) Standard Resistors

2 Film (commercial) Standard Resistors

2 Series/parallel (NBS) Resistance Boxes

2 Film (commercial) Standard Resistors

1 Series/parallel (NBS) Resistance Boxes

1 Series/parallel (NBS) Resistance Box

1 Series/parallel (NBS) Resistance Box Scott (NBS) Capacitive Discharge Apparatus

Scott (NBS) Capacitive Discharge Apparatus

Scott (NBS) Capacitive Discharge Apparatus

Scott (NBS) Capacitive Discharge Apparatus

Scott (NBS) Capacitive Discharge Apparatus

(MAP)

(MAP)

Primary Standards for Resistance at High Currents

10" 5 ohm

10

10"

10"

10

L

-4

1 Wolfe (commercial) High Current

1 Reichsanstalt (commercial) High

1 Wolfe (commercial) High Current

1 Reichsanstalt (commercial) High

1 Wolfe (commercial) High Current

1 Reichsanstalt (commercial) High

3 Reichsanstalt (commercial) High

3 Reichsanstalt (commercial) High

2 Reichsanstalt (commercial) High

1 Thomas (NBS) Standard Resistor

Standard

Current

Standard

Current

Standard

Current

Current

Current

Current

Standard

Standard

Standard

Standards

Standards

Standards

All except Primary Low Current Standards are also Working Standards

89
Sec. 2. PRECISION APPARATUS

Primary Standards

Primary Low Current Resistance Standards (Sec. 1) 2 NBS-speci f ied (commercial) Universal Ratio Sets

1 Dunfee-Dziuba (NBS) Primary Volt Ratio Standard

Two-stage (NBS) Inductive Voltage Divider

One-stage (commercial) Inductive Voltage Divider

Working Standards

2 NBS-specified (commercial) Universal Ratio Sets (same as above)

1 Dunfee-Dziuba (NBS) Primary Volt Ratio Standard (same as above)

2 Dziuba (NBS) Portable Volt Ratio Standards

1 Wheatstone (commercial) Bridge

1 Direct Reading Ratio Set (commercial)

1 Direct Reading Ratio Set (NBS)

12 Rosa (commercial) 100-k ohm Resistors

1 Two-stage (NBS) Inductive Voltage Divider (same as above)

1 One-stage (commercial) Inductive Voltage Divider (same as above)

Sec. 3. IMPEDANCE MEASUREMENTS

Primary Standards

1 Cutkosky (NBS) Valculable Capacitor

10 Cutkosky Fused-Silica Dielectric Capacitors (10 pF)

Primary Low Current Resistance Standards (See Sec. 1) Working Standards for Measurements of Capacitance

1 Cutkosky (NBS) Fused-Silica 10-pF Capacitor

1 Type-12 (commercial) Capacitance Bridge

5 Nitrogen Dielectric (commercial) Standard Capacitors

Working Standard for Measurement of Inductance

1 Maxwell-Wein (NBS) Bridge

Sec. 4. VOLTAGE

Primary Standard

Denenstein-Finnegan (NBS) AC Josephson Effect Apparatus

Working Standards

2 Cutkosky (NBS) Standard Cell Enclosures

1 NBS Oil Bath (NBS 11)

8 (commercial) Standard Cell Enclosures (MAP)

Sec. 5. AC-DC TRANSFER STANDARDS

Primary Standards

12 NBS-specified (commercial) Single-junction Evanohm Thermoelements

2 NBS-specified (commercial) Multi-junction Evanohm Thermoelements

Working Standards

2 sets (NBS) Coaxial Voltage Converters

2 sets NBS-specified (commercial) Thermal Current Converters

1 (NBS) Wattmeter

1 Yokagawa (commercial) Wattmeter

90
Sec. 6. TRANSFORMERS

Vol tage

Primary & Working Standards

Petersons (NBS) High Voltage Current Comparator Bridge

Current

Primary Standards

2 Souders (NBS) Two-stage Transformers

Working Standards

3 Compensate'd (NBS) Current Comparators

Sec. 7. HIGH VOLTAGE & ENERGY

DC Voltage Dividers

Primary Standards

Primary Resistance Standards (Sec. 1)

3 Park (NBS) High Voltage Dividers

Working Standards

3 Park NBS High Voltage Dividers (same as above)

10 Rosa (commercial) Standard Resistors

DC Kilovoltmeters

Primary Standards

Primary Resistance Standards (Sec. 1) Primary Voltage Standards (Sec. 4)

3 Park (NBS) High Voltage Dividers

Working Standards

3 Park (NBS) High Voltage Dividers (same as above)

1 (commercial) Digital Voltmeter

AC Voltage Dividers

Primary & Working Standards

Petersons (NBS) High Voltage Current Comparator Bridge

AC Kilovoltmeters

Primary Standards

Primary Voltage Standards (Sec. 4)

Petersons (NBS) High Voltage Current Comparator Bridge

Working Standards

Petersons (NBS) High Voltage Current Comparator Bridge (same as above)

1 (commercial) Electrostatic Voltmeter

91
Sec. 7. HIGH VOLTAGE & ENERGY (Continued;

Capacitors (High Voltage)

Primary Standards

Primary Capacitance Standard (Sec. 3)

Hill house (NBS) High Voltage Capacitor

Working Standards

2 Nitrogen Dielectric (commercial) Standard Capacitors

Watthour Meters

Primary Standards

Primary Voltage Standards (Sec. 4)

Primary Resistance Standards (Sec. 1)

Primary Time Interval Standards

Working Standards

Lentner (NBS) Current Comparator Energy System

4 Rotating (commercial) Watthour Meters

Impulse Dividers

Primary Standards

Primary Resistance Standards (Sec. 1)

Primary Voltage Standard Resistive Divider (Sec. 4)

Working Standard

Kerr Cell Apparatus

92
APPENDIX C

"CRITICAL ELECTRICAL MEASUREMENT NEEDS AND STANDARDS FOR MODERN ELECTRONIC INSTRUMENTATION"

WORKSHOP REPORT ABSTRACT, PREFACE, AND LIST OF ATTENDEES

Abstract

Recognizing the proliferation of sophisticated modern electronic instrumentation in the field of electrical measurements, the

Electricity Division of the National Bureau

of Standards recently initiated a new prog- ram in the general area of dynamic measurements and standards in support of such instrumentation. Recognizing further that the

vastness and complexity of the field would

require, at the earliest stages of the program, identification of the most critical

problem areas, the Electricity Division held

a workshop on 23 and 24 September 1974, at the Bureau's Gaithersburg site, to assist it

in ascertaining just what these areas in fact

were. The basic idea of the Workshop was to bring together a broadly representative group

of some 25 leading manufacturers and prime

users, working in a free and open atmosphere,

in order to have them delineate the present

and future critical support needs in the

field of dynamic electrical measurements for

modern electronic instrumentation, with em- phasis on physical standards, standardized

measurement methods, new calibration and

measurement assurance services, relevant

data, and, most important, new measurement

methodologies. The overall objectives of

the Workshop were generally met, and a num- ber of significant specific programs and

projects consistent with the mission of the

Electricity Division were identified. Three

categories broadly cover the needs as defined

at the Workshop:

1. The need to introduce time as a measurement parameter has resulted from the re- quirements of automatic test and control

systems. Specific areas considered critical

are:

a. Pulsed component measurements.

b. Dynamic performance characterization

for modern signal conditioning and data con- version devices: Digital-to-analog and

analog-to-digital converters, sample-andhold amplifiers, comparators, etc. Required

measurements include settling, aperture and

acquisition times. Basic new capabilities

will be needed, including precision, non- sinusoidal waveform generation and "standard"

digital-to-analog converters.

c. Methodologies and techniques for

characterizing precision ac and dc sources

and measurement devices with respect to settling time.

2. The emergence of measurements into the

"real world" or the production line via the

automatic system has introduced a host of new parameters. Areas requiring significant

effort include:

a. Investigation of transportable standards for val idation of static and dynamic

system performance: ac and dc voltage, impedance, pulses, settling times.

b. Techniques for characterizing sources

and measurement devices with respect to switched or dynamic loading.

c. Prediction of long-term performance,

reliability and lifetime from short-term

evaluation for a host of passive components,

semiconductors, and signal sources.

d. Noise standards and methodologies.

e. Continuing effort to improve trans- portable dc standards and dissemination at higher voltage levels.

f. Characterization of the ac line voltage-waveforms, under- and over-voltages,

transients, dynamic impedance, etc.

g. Inclusion of a capability for environmental control and variation for devices

and instruments under calibration, in all work undertaken by the Bureau.

h. Extension of Measurement Assurance

Programs (MAP's), especially into the above

new areas.

3. Recent work at the leading edge of mea- surement technology in electronics has

resulted in the need for new, or higher

accuracy, measurements. For example:

a. Increased dependence on capacitors as storage devices has made dielectric hysteresis a parameter of importance. Significant

measurement work should be started in this area.

b. Phase difference measurements to ex- tremely high accuracies are needed to calibrate industrial instrumentation.

c. Higher accuracy power measurements,

particularly for non-sinusoidal waveforms,

are needed to calibrate instrumentation in

the field.

93
d. Non-sinusoidal, high crest-factor

precision signals are needed to calibrate

true RMS converters and meters.

e. High accuracy electronic instrumentation requires reduction in ac calibration

uncertainties to 10 ppm, at least over the

audio range.

Longer-term, "frontier" areas in each of

the above groups were also identified. They

are not specified in the above listings since

they involve work at still more basic levels,

on ground that is presently terra incognita.

Nevertheless, they furnish insight into the

overall technical/philosophic climate, and

are useful for that reason. Items in this

category include:

1. Measurement problems in systems, including effects of history, measurement interactions, self-calibration, etc.

2. Evaluation and standardization of system

software.

3. Calibration of medical equipment.

4. Replacement of standard cells.

5. Replacement of thermal ac/dc converters.

6. Non-swept but broad-response measurements

-'impulse, noise, other.

7. Basic work on the physics of dielectric

absorption; component drift; device noise;

failure mechanisms; switches; and relays.

8. Provision of improved measurement accuracy for virtually all dynamic measurements,

at the wafer probe level.

In addition to the identification of these

specific and general needs, the Workshop provided strong evidence that both the manufacturing and user communities in the electronic

instrumentation field look to the Electricity

Division to provide the technological leadership in these newer areas that it has previously furnished in the more traditional

electrical measurement areas. Indeed, there

was a good deal of support for the idea of

keeping the electronics community informed

of Division work-in-progress, with emphasis

on projects targeted by the Workshop.

Keywords: Data conversion; dynamic measurements; electrical measurements;

electronic instrumentation; signal

conditioning; system; time domain.

Preface

It is apparent that the continued, or- derly growth of the electronics industry is increasingly hampered by a lack of standards

and methodologies in a number of crucially

important electrical measurement areas. The explosive growth of electronics during the last decade has brought with it a host of unsolved measurement problems. Yet at this critical historical juncture, electronics is a key technology in the prospective solutions to some of society's major

outstanding concerns including energy,

pollution, and productivity, to name but

three. It is, therefore, mandatory that a responsible, capable organization without

specific industrial affiliation, assume a leadership role in providing the necessary

measurement standards and methodologies, as well as in establishing new electrical mea- surement programs for which need is both

broad and identified, but whose implementation by specific industrial organizations

would be impractical.

The Electricity Division of the National

Bureau of Standards has traditionally dealt

with the measurement and generation of electrical signals in the dc to 30 kHz range, as well as with the properties of electrical

components in that range. In effect, with

the one possible minor (philosophical) ex- ception of phase, the Division has devoted

itself to amp! i tude or magni tude measurements for the various electrical parameters.

Some of these have been generated and/or

transmuted by mechanical and electro-mechanical means, others more recently by electronic, and others in future undoubtedly by

light, but the electrical (usually amplitude)

parameters have had similar properties and

required the same disciplines for measurement.

Now, automatic systems for test, process

control, medical applications, and so forth

are using electrical quantities at higher

speeds than heretofore. Ten years ago a precision dc, dialable 6- or 7-digit source did

not need a settling time specification; to- day, "programmable" sources have such specifications, since the system architect must

know how long it will take for this stimulus

to settle to within a given accuracy. In many instances total system speed, hence

all-important throughput, dollar payback

and/or justification, will be limited by

such settling times.

Thus, to precision electrical measurements

has been added the temporal parameter. In a manner analogous to the changeover from frequency to time domain work in filter synthesis in the early sixties, measurement must

now turn from exclusive attention in the

94
frequency domain, to characterization involving time as well, e.g., settling time to x microvolts in y microseconds is an electrical, analog, amplitude measurement with the

time factor added.

The Workshop which this report describes

and summarizes, is the means chosen by the

Electricity Division to establish direct

communication with key elements of the electronics industry among both manufacturers

and users, for the purpose of ascertaining

the industry's electrical measurement needs,

both general and specific. The enthusiasm

engendered by the Workshop itself attests to the fact that the time has come for leadership and action: In spite of difficult times

and severe cost pressures, 24 outstanding

organizations were willing to send top per- sonnel to the 2-day Workshop, disrupting

busy schedules, and at considerable financial

expense.

This report on the Workshop results must

inevitably reflect some of the author's technical biases, developed over the course of

25 years of varied design, management and

consulting experience in the fields of electronics, measurements and standards. During

that interval the technology has made tre- mendous advances, so that in many areas it

is virtually unrecognizable in comparison

with its major jumping-off point directly

following the Second World War. Indeed,

some of those advances are the direct result

of work accomplished at or backed up by programs of the Electricity Division of the NBS,

notably during what might be termed the mea- surement "explosion" of the decade of the

sixties.

By the same token, it is hoped that the

author's general breadth of design and mea- surement exposure during those same years of

experience, plus long-standing and continuing interactions with the Bureau, have provided a balanced view of the Workshop results

in this report. If this has been accomplished, the Division programs resulting

from the Workshop can form the basis for new

and crucially-important electrical measurement technology for electronics in the forthcoming decade.

It is impossible to convey here the ex- citement felt by virtually all participants

during the Workshop which helped to make the

author's task as moderator both rewarding

and enjoyable. The excitement was undoubtedly due in part to the professional challenge offered by the opportunity for open,

free-swinging exchange among the outstanding

group of participating engineers and scientists. But it clearly reflected, as well,

the participants' recognition of the large

and important work that so critically needs

to be done, and their enthusiastic reception

of the role of the Electricity Division in grasping the initiative to establish relevant new programs on a priority basis.

There was, in short, the excitement result

ing from the pervasive feeling of being part

of an undertaking that was both significant

and effective.

I should like to express my thanks to the Electricity Division and to Dr. Barry N. Taylor in particular, for giving me the opportunity to contribute to this important

work. I should also like to add my own personal thanks to the participants and their

organizations, as it is they who are truly

responsible for the Workshop's success.

Peter Richman

Consulting Engineer

95
WORKSHOP

Mr. Loyle E. Baltz

Principal Test Engineer

Ford Motor Company

Allen Park, MI

Mr. Eduard F. Boeckmann

Chief Engineer

Angstrohm Precision, Inc. Hagerstown, MD

Mr. Warren C. Collier

Manager, TM 500 Design, Engineering

Tektronix, Inc. Beaverton, OR

Mr. Thomas J. Coughlin

Product Engineering Manager

Component Systems

General Radio Company

Concord, MA

Mr. Jack D. Dougherty

Chief, DC & Low Frequency Standards

Laboratory

Newark Air Force Station, OH

Mr. George W. Herron

Senior Engineer

Keithley Instruments, Inc. Cleveland, OH

Mr. Donald E. Lawrence

Project Manager

Hewlett-Packard Company

Sunnyvale, CA

Mr. David R. Ludwig

Vice President

Advanced Development and Planning

Teledyne Phil brick

Dedham, MA

Mr. Edward P. Mueller

Biomedical Engineer

Food and Drug Administration

Rockville, MD

Mr. William A. Plice

Senior Project Staff Engineer

Honeywell, Inc. St. Louis Park, MN

Mr. Robert C. Raybold

Head, Institute for Basic Standards

Automation Group

National Bureau of Standards

Washington, DC

Mr. Bruce A. Renz

Head, Systems Measurement Section

American Electric Power Service Corp. New York, NY

ATTENDEES

Mr. David B. Schneider

Calibration Product Specialist, ATE John Fluke Manufacturing Co., Inc. Seattle,

Mr. Frederick B. Seeley

Chief, Electromagnetics Engineering Branch

U. S. Army Metrology and Calibration Center

Redstone Arsenal , AL

Mr. Robert M. Shaw

Fellow Engineer

Westinghouse Electric Corporation

Baltimore, MD

Mr. Lewis R. Smith

Engineering Manager

Analog Devices, Inc. Norwood, MA

Mr. Douqlas C. Strain

President

Electro Scientific Industries, Inc. Portland, OR

Mr. Robert P. Talambiras

Corporate Director of Advanced Development

Analogic Corporation

Elmwood Park, NJ

Mr. David S. Terrett

General Manager, Research & Development

ADT Security Systems

New York, NY

Mr. James C. Tripp

Engineering Supervisor

Linear Test Development Group

National Semiconductor Corporation

Santa Clara, CA

Mr. Robert H. Verity

Manager, Standards Laboratory

Leeds & Northrup Corporation

NorthWales, PA

Mr. William R. Walters

Chief Engineer

Codi Semiconductors

Fairlawn, NJ

Mr. F. Mansfield Young

Director of Engineering

Teradyne, Inc. Boston, MA

Mr. Ray A. Zuck

Director of Operations

Geometric Data Corporation

Wayne, PA

96
APPENDIX D

CODE: Type of Service

STANDARDS AND CALIBRATION LABORATORIES

S - Primary standards

C - Calibrations

B - Both

T - Testing

Type of Organization: C - Corporate laboratory

I - Instrument manufacturing

P - Calibration & metrology lab solely or mostly

No code indicates exact nature of service not known to authors.

Laboratory

Member

NCSL

Type of

Service

Code

Type of

Organization

Code

AEL, Colmar, PA AEL Communications, Lansdale, PA B C AIL Technological, City of Industry, CA B P AVCO Corporation, Charleston, SC X B C AVCO Corporation, Lycoming Div., Stratford, CT C C AVCO Corporation, Richmond, IN c C Aerojet Nuclear, Idaho Falls, ID X B c Ailtech Cutler Hammer Co., Farmingdale, NY C c Airtron Div., Litton Systems, Morris Plains, NY C c AIT Systems, Morristown, NJ

All is Chalmers Mfg. Co., Milwaukee, WI X B c American Electronic Labs, Lansdale, PA B p American Geophysical & Instrumentation, Gardena, CA B p American Instrument Service, Philadelphia, PA B p Ark Electronics, Willow Grove, PA C p B&K Instruments, Cleveland, OH C p BLH Electronics, Waltham, MA ? Ballantine Laboratories, Boonton, NJ C i Ball Brothers Research, Boulder, CO X B c Bath Iron Works Corporation, Bath, ME X C c Beckman Instruments, Fullerton, CA X B c Beech Aircraft Corporation, Wichita, KS X B c Bell Aerospace Co., Buffalo, NY X B c Bendix Corporation, Teterboro, NJ X S c James G. Biddle Co., Plymouth Meeting, PA X B I Blake Electronics, Inc., Greendale, WI C Boeing, Wichita, KS C c Boeing Aerospace Group, Seattle, WA X B c Brunswick Engineering, Inc., Northbrook, IL C Burroughs Corporation, Paoli, PA B c Calibration Central, Alexandria, VA C C Certified Calibration Labs, Inc., Philadelphia, PA S C p Collins Radio Co., Cedar Rapids, IA X B c Communications Satellite Corp., Clarksburg, MD X B c Computer Diode Co., Fairlawn, NJ X B I

Sources: A Directory of Standards Laboratories, 1974 Edition, National Council of

Standards Laboratories, Washington, DC, 1974.

Electronics '75 Buyers' Guide, McGraw-Hill, New York, NY, 1975. eem 75/76, electronic engineers master catalog, United Technical Publications,

Garden City, NY, 1975.

Electronic Design's Gold Book, Hayden Publishing Company, Rochelle Park, NJ, 1975.

97
Laboratory

Member

NCSL

Type of

Service

Code

Type of

Organization

Code

Comtel Corporation, Detroit, MI Cox & Co., New York, NY DBA Systems, Melbourne, FL Dal mo Victor, Belmont, CA Davidson Optronics, West Covina, CA Dayton T. Brown, Inc., Bohemia, NY Detroit Edison Co., Detroit, MI Dillon, W. C. & Co., Van Nuys, CA Douglas Instrument Lab, Boston, MA

Dow Chemical Co., Rocky Flats, CO EG&G, Albuquerque, NM EG&G, Las Vegas, NV

E. I. DuPont de Nemours, Aiken, SC EIL Instruments, Timonium, MD Electrical Instrument Service, Mt. Vernon, NY Electrical Science Services, Cambridge, MA Electrical Testing Labs, Inc., New York, NY Electro Scientific Industries, Portland, OR Electronic Marketing Associates, Kensington, MD Eppley Laboratories, Newport, RI E-Systems, Dallas, TX Fairchild Space & Electronics, Germantown, MD Fischer/Porter Co., Warminister, PA John Fluke Mfg. Co., Seattle, WA Foxboro Co., Foxboro, MA Gage Corp., Philadelphia, PA General Dynamics, Groton, CT General Dynamics, Quincy, MA General Dynamics, San Diego, CA

General Electric, Bay St. Louis, MS General Electric, Dallas, TX General Electric, Pittsfield, MA General Electric, Schenectady, NY General Electric, Seattle, WA General Electric, Syracuse, NY General Electric Instrumentation, Inglewood, CA General Environments Corp., Morton Grove, IL GENRAD, West Concord, MA Greentron, Inc., Greenville, SC Guildline Instruments, Inc., Larchmont, NY H-B Instrument Co., Philadelphia, PA Hamilton Technology, Lancaster, PA Harris Intertype, Melbourne, FL Hercules Inc., Magna, UT Hewlett-Packard Co., Palo Alto, CA Hewlett-Packard, Santa Clara, CA Hoffman Engineering Corp., Old Greenwich, CT Honeywel

1

, Inc., Aerospace Div., St. Petersburg, FL Honeywel

1

, Inc., Annapolis, MD

Honeywel

1

, Inc., Burlington, MA Honeywel

1

, Inc., Chicago, IL Honeywel

1

, Inc., Cleveland, OH

Honeywel

1

, Inc., Denver, CO Honeywel

1

, Inc., Detroit, MI Honeywel

1

, Inc., Downey, CA Honeywel

1

, Inc., Hazelwood, MO Honeywel

1

, InCo, Fairborn, OH Honeywel

1

, Inc., King of Prussia, PA Honeywel

1

, Inc., Minneapolis, MI

c X B c

c

X B c

c P

X B P

X B c

B P

X c c

c c

X c c

x B c

X B p

X B p c

X B,T p

X B I c

B I X B c

c c

c

X B I X B c

X c p

X B c

c c

X B c

x B c

c c

x B c x B c

c c

x B c

s c

c p

x B I c p

x s I c I r c

x B o

xA c c

c I B I C

X C P

C P

C p

c p

c p

X B p

c p

c p

c p

c p

c p

B c

98
Laboratory

Member

NCSL

Type of

Service

Code

Type of

Organization

Code

Honeywell, Inc., San Diego, CA

Honeywell, Inc., Springfield, VA Honeywell, Inc., Sunnyvale, CA

Honeywell, Inc., Westfield, NJ Hy-Cal Engineering, Sante Fe Springs, CA

Hughes Aircraft, Culver City, CA

Hughes Aircraft, Fullerton, CA

I IT Research Institute, Chicago, IL Incal Service Corp., Stanford, CT Infinite, Inc., Cape Canaveral, FL Inland Testing Labs, Inc., Morton Grove, IL Instrument Labs Corp., Chicago, IL Intergalactic Corporation, Salt Lake City, UT Julie Research Labs, New York, NY

KC Calibration Lab, Kansas City, KS

Kahl Scientific Instr. Corp., El Cajon, CA

Keithley Instruments, Cleveland, OH

Key Instruments, Gardena, CA

LTI Research Foundation, Lowell, MA

Lab Instrument Service, Campbell, CA Lear Siegler, Santa Monica, CA

Leeds & Northrup, North Wales, PA Lehighton Electronics, Lehighton, PA Librascope, Div. of Litton, Glendale, CA

Lockheed Aircraft Service, Ontario, CA

Lockheed-California, Burbank, CA

Lockheed Electronics, Houston, TX Lockheed-Georgia, Marietta, GA

Lockheed Missiles & Space Co., Sunnyvale, CA

3M Corp., St. Paul , MN Main Electronics, Wichita, KS Mancib Co., Burlington, MA Martin Marietta, Denver, CO Martin Marietta, Marietta, GA

McDonnell Aircraft Co., St. Louis, MO

McDonnell Douglas Astronautics, Long Beach, CA

McDonnell Douglas Astronautics,

West Huntington Beach, CA

Micro-Radian . Instruments, Seal Beach, CA

JR Miles Corporation, Elk Grove Village, IL Montedoro-Whitney, San Luis Obispo, CA

National Astro Labs, Burbank, CA National Camera, Inc., Englewood, CO New Era Products, Cleveland, OH Newport News Ship Building & Drydock, Newport News, VA New York Testing Labs, Westbury, NY North American Rockwell, Anaheim, CA North American Rockwell, Rocketdyne Div., McGregor, TX Nucleus Corp., Madison Heights, MI Ogden Technology Labs, Fullerton, CA Optronic Labs, Silver Spring, MD Ormond Inc., Sante Fe Springs, CA PTS Electronics, Inc., Bloomington, IN Penn Airborne Products, Southampton, PA Pioneer/Instrumentation, Cleveland, OH QVS Inc., East Orange, NJ RCA, Gov't Services, Moorestown, NJ RCA International Service Corp., Camden, NJ RCA International Service Corp., Patrick AFB, FL

c c

c p c P

c P

s P B c c c

x B p

C P C P x s p

c p c

B I c

s p B I

xA

c

B

pp c

c c

x B

c

I

C c C c

Ay R r

Ax B c

Rur X B c

X B c

r<-> p r p Ax B c

xA B c

YA B c

A R r

Ax B c

c p

rB p B

r

p

r P

r r

c p

X B c

B c

C p B p

C

C p CB I B p C

B c B c

X B c

99
Laboratory

Member

NCSL

Type of

Service

Code

Type of

Organization

Code

Ram Meter, Inc., Ferndale, MI c P Ramsey Engineering, Anaheim, CA C P Raytheon Submarine Signal, Portsmouth, RI X B C Reliance Electric Co., Cleveland, OH c P Robertshaw Controls Co., Richmond, VA C Rockwell Engineering Co., Indianapolis, IN c P Rutherford Research Products, Rutherford, NJ c P SSCO Standards Laboratory, Detroit, MI B P

/* _i n • i ftii ft 1 1 i Sanders Associates, Nashua, NH X B C Scientific Leasing Services, Lake Success, NY c Sensitive Research Instruments, Mt. Vernon, NY c P Simpson Electric Co., Elgin, IL c Singer Co., Kearfott Div., Little Falls, ND X B c Sperry Electronic Tube Div., Gainesville, FL B c Stabro Labs, Salt Lake City, UT B p States Electronic Corp., Blundworth Marine, Linden, NJ C c Sunshine Scientific Instruments, Inc., Philadelphia , PA B p Sweetman Calibration Services, Pennsauken, NJ X B p Systron Donner, Concord, CA X S c TRW Systems, Redondo Beach, LA X B c Taylor Instrument, Process Control Div., Rochester, NY B c Technicians Unlimited, Lexington, MA C Teledyne Brown Engineering, Huntsville, AL c c Teledyne McCormick Selph, Hollister, CA c c Teledyne Systems Co., Northridge, CA X B c Texas Instruments, Houston, TX C p Thwing-Albert Instrument, Philadelphia, PA c p Transducers, Whittier, CA

UN I VAC, Utica, NY X S

' c Union Carbide, Oak Ridge, TN X B c Varian, Palo Alto, CA X S I Viking Labs, Sunnyvale, CA B c Westinghouse Electric Co., Baltimore, MD X B c Westinghouse Electric Co., Pittsburgh, PA X B c Weston Instruments, Newark, NJ X B I

This listing of calibration and standards laboratories is not to be taken as an endorsement

by the National Bureau of Standards of laboratories included, or as an indication of the

quality of calibrations available from either listed or unlisted organizations.

100
APPENDIX E

TYPICAL MEASUREMENT SUPPORT STRUCTURES

On the following pages are representative functional diagrams of two corporations and

their measurement support structures. On the following page is depicted the structure by

which the U.S. Air Force, through a network of over 140 calibration facilities, provides

world wide measurement support for over $400 million worth of electronic test equipment

with which maintenance and operational support for weapons, communications, navigation,

guidance and medical systems are supplied.

101
c

(J Eo

CD <D

SO.

r—

O

O r- f- *-» o to

OJ J3 Q. L- E >- cu o

cr >— cr. O TO ft) <_) t_> oO _J

A

A ^ t o i-

•t- CO

T3 4-> 1- i- O O

O. CL

V- Q. O 3

=3U

QJ S4-> -t-> 03 CO

S_o +-> CL Ss- o

CL

CL CO

(T3 U -!-> r- C

CL O) >> E I— o>

s-Cl

co(O0)

102
03

t- 4JU

3

i~

4-> "5 on

S~O 4J Q. i. %. O

O D.CL

reu

C

Q. <D >) E

h- ^

1/5

OJ

->- —> >—

ooa:

a o-r- Q

103
Measurement Tools from Instrument Industry

Federal Government

Legal Requirements

Naval Observatory

Washington, DC

Measurement Services from NBS Electricity Division - 211.00 Mechanica Division - 213.00

Heat Division - 221 .00 Optical Physics Division - 232.00 Electromagnetics Division - 272.00 Time and Frequency Division - 273.00

Principles of Physics and Technical Literature

AIR FORCE MEASUREMENT

STANDARDS LABORATORY

(Operating Budget $4,465,000)

Army Standards Laboratory

Redstone, AL

>

Over 140 Preci Equipment L< (Operating Budg<

;ion Measurement

iboratories

;t - $43,000,000)

±

Foreign Government Laboratories

Ethopia Korea Greece Malaysia

India Nat. China Indonesia Phillipine Islands

Iran Spain

Israel Thailand

Jordon Turkey

Maintenance and Repair Depots

I

iL.

All Aircraft, Missile & Ground Test Equipment

1

Other U.S. Government,

Army, Navy, FAA Test Equipment

TOTAL AIR FORCE BUDGET FOR FY 73: $23,737,000,000

Air Force Metrology Structure - FY 73

104
REFERENCES

"1] The International System of Units (SI), Nat. Bur. Stand. (U.S.), Spec. Publ . 330,

4

Q pages (1972).

"2] Thomas, J. A., Peterson, C, Cooter, I. E., and Kotter, F. R. , An Absolute Measurement

of Resistance by the Wenner Method, J. Res. NBS (U.S.), 43, No. 4 (1949) RP2025.

"3] Cutkosky, R. D., New NMS Measurements of the Absolute Farad and Ohm, IEEE Trans.

Instrum. Meas. IM-23 , No. 4, 305-309 (Dec. 1974).

"4] Cutkosky, R. D. and Lee, L. H., Improved Ten-picofarad Fused-silica Dielectric

Capacitor, J. Res. NBS (U.S.), 69C (Engr. & Instr.), No. 3, 173-179 (July-Sept. 1965).

"5] Field, B. F. , Finnegan, T. F., and Toots, J., Volt Maintenance at NBS via 2e/h: A New Definition of the NBS Volt, Metrologia 9, No. 4, 155-156 (1973).

'6] American National Standards Institute 1976 Catalog (Am. Nat. Stand. Inst., NY, 1974).

"7] Harvey, G. F., editor, Standard and Practices for Instrumentation - Fourth Edition

(Instr. Soc. Amer. , Pittsburgh, PA 1974).

"8] Electronic Engineers Master (United Technical Publications, Garden City, NY, 1974).

"9] U.S. Industrial 1975 Outlook with projections to 1980, Department of Commerce,

432 pages (1975).

10] U.S. Industrial 1976 Outlook with projections to 1985, Department of Commerce,

465 pages (1976).

"11] Evenson, K. M. , Wells, J. S., Peterson, F. R., Danielson, B. L., Day, G. W. , Barger,

R. L., and Hall, J. L., Speed of Light from Direct Frequency and Wavelength Measurements of the Methane-Stabilized Laser, Phys. Rev. Lett. 29, 1346 (1972).

'12] Cohen, E. R. and Taylor, B. N., The 1973 Least-squares Adjustment of the Fundamental

Constants, Journal of Physical and Chemical Reference Data, 2, No. 4, 663 (1973).

"13] Richman, P., editor, Critical Electrical Measurement Needs and Standards for Modern

Electronic Instrumentation, Nat. Bur. Stand. (U.S.), Tech. Note 865, 74 pages (May

1975) . "14] A Directory of Standards Laboratories - 1975 Edition (National Conference of Standards Laboratories, Washington, 1975).

'15] Use of off-the-shelf electronic test equipment to reduce costs, shorten leadtimes, assure reliability, and simplify logistics, Report of the Task Force on Science Board,

Department of Defense, Washington (Feb. 1976).

"16] Minutes of Automatic Test Equipment Conference and Workshop (April 1976) by Industry

Ad Hoc Automatic Test Equipment Project for the Navy, National Project for the Navy,

National Security Industrial Assn., Washington, DC, to be published late 1976.

17] Technical Manual - Calibration Technical Orders, Responsibilities and Calibration

Measurement Areas, TO 33K-1-100, U.S. Air Force (March 1972).

;i8] 1975 Annual Statistical Report, Electrical World 183, No. 6, 44-75 (March 1975).

'19] A Summary of Program Emphasis for 1976, Electric Power Research Institute, Palo Alto,

CA (1976).

"20] U.S. Industrial Outlook 1974 with projections to 1980, Department of Commerce,

385 pages (1974).

;21] U.S. Markets Forecast 1977, Electronics, Vol. 50, No. 1, p. 90 (Jan. 6, 1977).

'22] Houghton, S., Transfer of the Ki lowatthour, IEEE Trans, on Power Apparatus and Systems,

Vol. P94, No. 4, 1232 (Jul. /Aug. 1975).

"23] A National Plan for Energy Research, Development and Demonstration: Creating Energy

Choices for the Future, ERDA 48, Vol. 2, Energy Research and Development Agency

(June 1975).

'24] IEEE Power Engineering Society Proceedings of Conference on Research for the Power

Industry, IEEE, NY (1973).

^25] Underground Power Transmission, Electric Research Council, NY (1971).

'26] Electric Utilities Industry Research and Development Goals through the Year 2000,

Electric Research Council, NY (1971).

;27] EPRI Transmission and Distribution Division Report, EPRI Journal, Vol. 3, 28 (April 1976)

"28] EPRI Transmission and Distribution Division Report, EPRI Journal, Vol. 1, 40 (February

1976) . ;29] EPRI Transmission and Distribution Division Report, EPRI Journal, Vol. 2, 34 (March 1976)

"30] Allen, R., Instrumentation: smaller and smarter, IEEE Spectrum, Vol. 13, No. 1, 68

(Jan 1976).

[31] Eicke, W.G., Jr., and Auxier, L. M. , Regional Maintenance of the Volt Using NBS Volt

Transfer Techniques, IEEE Trans. Instrum. Meas., IM-23 , No. 4, 290-294 (Dec. 1974).

105
BIBLIOGRAPHY

1. Cochran, R. D., Measures for Progress, Nat. Bur. Stand. (U.S. ) Misc. Publ . 275,

703 pages (1966).

2. Coombs, C. F. Jr., Editor, Basic Electronic Instrument Handbook (McGraw-Hill,

New York, NY, 1972).

3. Department of Commerce, Current Industrial Reports: Selected Instruments and Related

Products (U.S. Department of Commerce, Washington, DC, 28 pages, 1973).

4. Driscoll, R. L., Measurement of Current with a Pellat-type Electrodynamometer, J. Res. NBS (U.S.), 60, No. 4, 287-296 (April 1958).

5. Driscoll, R. L. and Cutkosky, R. D., Measurement of Current with the National Bureau

cf Standards Current Balance, J. Res. NBS (U.S.), 60, No. 4, 297-305 (April 1958).

6. Dun and Bradstreet, Million Dollar Directory, 1974, (New York, NY, 1974).

7. Harris, F. K. , Electrical Measurements (John Wiley & Sons, New York, NY, 784 pages,

1952).

8. Hermach, F. L. and Dziuba, R. F., Editors, Precision Measurement and Calibration - Electricity - Low Frequency, NBS (U.S.) Spec. Publ. 300, Vol. 3, 489 pages (Dec. 1968).

9. Lerner, W., Statistical Abstract of the U.S. - 1975, (Department of Commerce, Washington,

DC, 1028 pages, 1974).

10. Office of Management and Budget (U.S.), Standard Industrial Classification Manual,

1972 (U.S. Government Printing Office, 150 pages, 1972).

11. Olsen, P. T. and Williams, W. E., A More Accurate Determination of y' Through Im- proved Techniques, IEEE Trans. Instrum. Meas., IM-23 , No. 4, 302-305p (Dec. 1974).

12. Perry, John, The Story of Standards, (Funk & Wagnells, New York, NY, 271 pages, 1955).

13. Terman, F. E. and Pettit, J. M., Electronic Measurements, 2nd Edition (McGraw-Hill,

New York, NY 707 pages, 1952).

106
INSTRUMENT SOCIETY of AMERICA

22nd Annual ISA Conference and Exhibit • September 11-14. 1967, Chicago. Illinois

Preprint

Number MS-l-MESTIND-67

A 11 righta rmmd For pcrmiuion to publish, u/ritt ISA. 530 Win. P.iui Plan. Putthurlh. Pa. 16319

LOW-FREQUENCY ELECTRICAL CALIBRATIONS AT THE NATIONAL BUREAU OF STANDARDS

F. L. Hermach - Senior Mtn-iber

Chief, Electrical Instruments Section

National Bureau of Standards

Washington, D. C.

ABSTRACT

Chard are presented to show the present range and accuracy of NBS calibrations of standards of resistance, capacitance, inductance, voltage, and

current, from direct current through 50 kHz. The

chains of measurements by which these and other

calibrations are related to the basic NBS standards

of voltage and resistance are also shown.

INTRODUCTION

There are many electric and magnetic quantities

to be measured, such as resistance, voltage and flux

density. Measurements of most of them are made over many decades of both magnitude and frequency, with

an amazing variety of standards and instruments.

Thus it is certainly impractical to offer a calibration service at NBS to cover all of these variables. NBS concentrates instead on the calibration*

of a few types of standards of the highest stability

and accuracy. Because of this stability, the scientist or engineer can then verify the accuracy of

his measurements with only infrequent periodic

calibrations of these standards.

This paper briefly describes these calibration

services and the NBS electrical standards on which

they are based. It is a revision of an earlier

paper 1, with charts which now show the accuracies

normally available in calibrations, at established

fees, of commercial standards of the highest grade?.

(This information was not given in the earlier

paper.) These charts also show the best accuracies

attainable at present in the calibration of "ideal"

standards, by special tests with the NBS research

equipment. This equipment is generally not available for normal calibration services.

NBS STANDARDS

Figure 1 shows the major electrical standards

which are used by NBS in the calibration program.

The lines indicate the major relationships between

them. For clarity, some minor relationships and

the kinds of calibrations performed are now shown.

^Calibration is used here as the process of making appropriate measurements to determine the correct value of a standard.

The internationally-accepted prototype standards of mass, length, and time are given in the top row of the figure. The meter is now defined as a certain number of wave lengths of the orange-red

line of Krypton 86 and the meter bar is a reference,

rather than a prototype, standard-*. The atomic

second is now defined in terms of a specified

transition of Cesium 133, and stable oscillators

serve as working standards of frequency (Hz)^.

The kilogram is still defined as the mass of the

prototype Pt-Ir standard.

Two experiments are performed at NBS to determine the basic electrical units In terms of these

three standards and two measured constants, the speed of light in vacuo (c) and the acceleration

of gravity (g) . They are simple in principle but

extremely difficult and involved in practice be- cause of the accuracy required. One experiment

consists in constructing a capacitor from gage bars

and computing its capacitance in electromagnetic

units (about 1 pF) from the length of the bars and

the speed of light. With suitable bridges the step up is made at 1592 Hz to two 10,000-pF capacitors (Xc 10,000 Q), across to two 10,000-Q

resistors of known ac-dc difference, and down to 1-w resistors^. The final step Is taken because

l-ll Thomas-type resistors are still the most stable

Impedance standards known. The average resistance

of a group of such resistors serves to maintain the ohm at NBS between such absolute dstezm inations

The other experiment consists in "weighing the

ampere" with a current balance, and is thus based

directly on the defintion of th* ampere in terms of

the force between current- carrying conductors^.

One conductor (coil) is suspended from one arm of a balance, and the change in fores when its current

Is reversed is measured in terms of the acceleration of gravity on a known mass. The voltage drop

In a 1-Q standard resistor carrying this measured

direct current is then used to determine the emf

of a group of saturated standard cells, which, in

turn, maintain the volt at NBS.

Direct- reading ratio-sets were developed st

NBS to calibrate resistors by the substitution

method, and to step up and down on the resistance

scale 7

. No line leads to them in Figure 1 because

their sccuracy depends on ratios of resistors, not

on the unit of resistance. Standard cells are calibrated by connecting the known and unknown cells in opposition and measuring the small voltage difference with a low-range thermofree potentiometer.

Potentiometers and volt boxes are calibrated with

Superior masher* refer to' similarly-mmnbersd references at the end of this papsr.

107
universal and direct-reading rat to-sets ' »° . With

these the user can then extend the d-c voltage

scale very accurately.

AC-DC transfer Instruments are comparators for

determining the equality of a-c and d-c currents,

a-c and d-c voltages, and a-c and d-c powers^.lO.

At NBS they serve chiefly to determine how well

other ac-dc comparators do this. Since such com- parators are very stable, the user can then make

accurate a-c measurements based on his known d-c

s tandards

.

A-C bridges extend the scale of capacitance

and Inductance measurements, but NBS calibrates

only the more stable capacitors and Inductors, with

which bridges can be checked. A-C voltage dividers

or a-c potentiometric networks are used to determine

the ratios (expressed as complex numbers) of the

phasor voltages and currents of the NBS standard

transformers and inductive voltage dividers, which,

in turn, are used to calibrate other standards.

In principle the ratios of these networks are also

independent of units, but in practice some of the component impedors are measured with the d-c ratio

sets and a-c bridges. Build-up or "boot-strap"

techniques are used for the newer current comparators and audio frequency current transformers^*

The standard watthour meters and standards for magnetic measurements depend on the other standards

as shown in Figure 113.1*.

ACCURACY CHARTS

The charts of figures 2 through 6 show the uncertainty (accuracy) in the calibrations of

standards at NBS, as a function of the magnitude

of the quantity being measured. Because of the

wide ranges, logarithmic scales are used. In each

figure the uncertainty is considered as the "limit

of error" which should be exceeded only very rarely. It includes the calculated imprecision (at

least three times the standard deviation of the average) plus the estimated limits of the systematic errors of the NBS working standards and the calibration process. It does not include the differences between the absolute units and the legal

units maintained at NBS 5 » 6.

The solid lines and circles in each figure

show the accuracies normally available in the

calibration of commercially available standards of

the highest grade, by established techniques, and

for the fees given in the published NBS fee schedules. These accuracies include allowances for

short-time variations in the standards under test under the controlled conditions at NBS, but not for

possible larger variations in other environments nor

for long-time drifts or other changes. The dotted

lines and open circles show the best accuracies

that might be attainable by special techniques,

and, with repeated tests made with great care and

effort in the calibration of "Ideal standards"

(which are perfectly stable and free from environmental influences).

Figure 2, for resistors, shows the familiar

"accuracy triangle," with the peak at the value of

the 1-ohm basic standard. Note that only decimal

multiples and sub-multiples of 1 ohm are shown; NBS does not usually offer calibrations of other values

nor does it maintain an accurate resistancemeasuring facility as such. However, stable standards of any nominal value can be calibrated at greater cost and lesser accuracy than standards of neighboring values on the decimal scale. Thomastype 1-ohm resistors are normally calibrated to

1 ppm in terms of the ohm maintained at NBS (i.e.,

the uncertainty is 1 ppm as shown by the solid

circle indicating available accuracy at this point).

High quality direct-reading and universal ratio sets

(not shown in Figure 2) can be calibrated with an uncertainty not exceeding 1 ppm of input resistance.

Using relatively simple equipment and established

calibration procedures many users of these ratio

devices could themselves perform these calibrations

with comparable uncertainties*.

The chart for capacitance measurements, Figure

3, should be three-dimensional, with frequency as another independent variable. One would expect, in

a rough way, an "accuracy cone" with the peak at

1 pF and 1592 Hz, where the best measurements with

the computable capacitor have been made.

The dotted line in Figure 3 shows the sensitivity limit at 1592 Hz of a new research bridge

which is being developed In conjunction with a new calculable capacitor. The estimated accuracy of the capacitor is also shown. The accuracy of the

research bridge for stepping up and down from Its value (0.5 pF) will, of course, be somewhat poorer

than the sensitivity limit Indicated.

Some of the contours for normal calibrations of

the best available commercial capacitors are shown

as solid lines in the figures. Note the extremely

wide ranges In magnitude (10~^ to 10^ ^F) and In frequency (60 to 10000 Hz).

Some sealed three- terminal air capacitors from

10 to 1000 pF have shown unusual stabilities. Pro- jected improvements in the NBS ratio arm bridge

should make it possible to measure such capacitors

to at least 10 ppm. Even better accuracy may be

required to calibrate 10 pF sealed capacitors of

fused silica with silvered electrodes, which are apparently stable to better than 1 ppm/year^.

Standard inductors are calibrated with standard

capacitors in a Maxwell Wien bridge. Some of the contours in the accuracy cone are shown in Figure 4.

*This is true for many types of range-extending

ratio standards (which do not involve the units).

In many cases, however, the equipment and techniques

are quite involved, and verification of the accuracy

of the results by "round-robin tests" of a traveling

standard in several laboratories, or by a direct

calibration of a few ranges at NBS, may be very

des lrable.

M5-1-MESTIND-67

108
The available accuracy li generally Halted by the

environmental conditions affecting the bridge and

the standards being calibrated.

The charts of voltage measurements (Figures 5 and 6) are complicated not only because of applied

frequency but also because NBS does not offer a voltage calibration service aa such. Only the d-c

voltage of a standard cell is available directly

(and is determined to 1 ppm in terms of the volt

maintained at NBS). The scale is extended in voltage and frequency by the calibrations of ratio

standards and ac-dc transfer standards. With these

and a standard cell the user can then make accurate

d-c and a-c measurements. The ratio standards

(potentiometers, fixed resistance dividers, and

transformers) are generally used with respect to some reference voltage, such as the emf of a cell

for d-c ratio standards, and 100 V (approximately

line voltage) for a-c standards. They have upper

and lower voltage limits. The chart therefore

shows the accuracy of the ratio V/V r , where V is given along the abscissa, and V r

is this reference.

Note that this is the accuracy of the indicated

ratio, not of a fraction of the applied voltage.

(A 1000/1 volt box certified to 0.001 percent of

ratio has one-hundredth the uncertainty of a 1-ppmof-input voltage divider at this same ratio.)

Potentiometers of the highest grade are normally

calibrated to 3 ppm at 1 V, volt boxes to 10 ppm

of ratio, decade transformers to 0.5 ppm of input

voltage at 1 kHz, and voltage transformers to 0.03

percent of ratio at 60 Hz. AC-DC transfer instru- ments for voltage measurements are normally calibrated to 50 ppm from 20 Hz to 20 kHz and 100 ppm

at 5 Hz to 50 kHz. The dotted lines show the "attainable" accuracies.

The chart for current measurements, Figure 7,

is similar. Here NBS offers only calibrations of

d-c shunts (for use with a potentiometer), a-c ratio standards, and ac-dc transfer standards. A

chart for power and energy measurements would be

based on the NBS standard transfer wattmeter, which

is known to better than 50 ppm of the applied

volt-amperes at 60 Hz and 0.1 percent at 2 kHz.

The NBS group of five standard watthour meters «re calibrated periodically in terms of the d-c standards through this wattmeter, and are in turn used

to calibrate standard watthour meters to 0.05

percent at 60 Hz.

SUMMARY

These charts show the accuracies normally

available a_t present , \a well as the best attainable under the rec Lrictiona given. The ranges and accuracies of low- frequency electrical measurements are increasing rapidly, so that this is a far- from-static field. The results of research

and development at NBS and elsewhere will be incorporated in improved NBS calibration services as rapidly as resources permit.

The author thanks Chester Peterson for Figures

2, 3, and 4, and for the ideas he has expressed

during discussions of this subject.

KEYWORDS

electrical calibrations, electrical standard*, lowfrequency calibrations, low-frequency standards.

REFERENCES

1. Low Frequency Electrical Calibrations at the National Bureau of Standards, F. L. Hermach, NBS Miscellaneous Publication 248, Paper No. 1.5, 1962.

2. Calibration and Test Services of the National Bureau of Standards, NBS Miscellaneous Publication

250, (consult latest edition).

3. Actions of 11th Conference on Weights and

Measures, NBS Technical News Bulletin, 44, 199, Dec. 1960.

4. World Sets Atomic Definition of Time, NBS Technical News Bulletin, 48, 209, Dec. 1964.

5. Evaluation of the NBS Unit of Resistance Based

on a Computable Capacitor, R. D. Cutkosky, Journal

of Research NBS, 65A, 147, 1961.

6. Measurement of Current with the NBS Current Balance, R. L. Driscoll and R. D. Cutkosky,

Journal of Research NBS, 60, 297, 1958.

7. Precision Resistors and Their Measurements, J.

L. Thomas, NBS Circular 470, 1948.

8. Method for Calibrating a Standard Volt Box,

B. L. Dunfee, Journal of Research NBS, 67C, 1963.

9. Thermal Converters for Audio-Frequency Voltage Measurements of High Accuracy, F. L. Hermach and

E. S. Williams, IEEE Trans, on Instrumentation and

Measurement, Vol. IM-15, No. 4, Dec. 1966.

10. Standard Electrodynamlc Wattmeter and AC-DC

Transfer Instrument, J. H. Park and A. B. Lewis,

Journal of Research NBS, Vol. 25, p. 545, 1940.

11. The Precise Measurement of Current Ratios, N. L. Kusters, IEEE Trans, on Instrumentation and

Measurement, Vol. IM-13, No. 4, Dec. 1964.

12. The Design and Performance of Multirange Cur- rent Transformer Standards for Audio Frequencies,

B. L. Dunfee, IEEE Trans, on Instrumentation and

Measurement, Vol. IM-14, No. 4, Dec. 1965.

13. Precise Comparison Method of Testing AC Watthour Meters, A. W. Splnks and T. L. Zapf, Journal

of Research NBS, 53, 95, 1954.

14. Basic Magnetic Quantities and the Measurement

of Magnetic Properties of Materials, R. L. Sanford

and I. L. Cooter, NBS Monograph 47, 1962.

15. Improved 10-Picofarad Fused Silica Dielectric

Capacitor, R. D. Cutkosky and L. H. Lee, Journal of

Research NBS, Sect. C. , Vol. 69C, No. 3, JulySept. 1965.

M5-1-MESTIND-67

109
ILLUSTRATIONS

Figure 1. NBS Electrical Standard!

Figure 2. Resistance Measurements

Figure 3. Capacitance Measurements

Figure 4. Inductance Measurements

Figure 5. D-C Voltage and Voltage-Ratio Measurements

Note! The accuracy of ratio V/Vr is shown for ratio standards, with

V r - 1 V.

Figure 6. A-C Voltage and Voltage-Ratio Measurements

Note: The accuracy of ratio V/Vr

is shown for ratio standards, with

V r - 100 V.

Figure 7. Current and Current-Ratio Measurements

Notei The accuracy of ratio I/Ir

is shown for ratio standards, with

I - 5 A.

110
FIGURE 1

IT"

Wavelength

r 133

Ca Time Interval Kllogra

Ccatpu table Capacitor

Standard Capacitors

& Bridge

Standard

Inductors

J. Bridge

Magnetic

Standards

Oscillator & Frequency Dividers

Standard Resistors

AC Dividers

& Potentiometer

Networks

Standard Transformers

DC Ratio

Sets

AC DC Transfer

Standards

V, I, P

Standard DC

Voltage Divider

Currant

Balance

Standard

Cells

DC Potentlosu ter

Standard Watthour

Maters

FIGURE 2

O

Q.

s

O

>>c

'o

CDo

c

=)

10'

10'

10

s

10*

I0

J

10*

1 1 1

o

o o

o

- ° o

CP/ TEW- 12/20/66

<JJ£s>

° •

• O •

•

- K

i

via Wenner Bridge O via Wheatstone

~0• 0

Bridae

O

• • •

via Capacitance Discharge kJ 1 I

i I

H

PPM

0.1

10

100

0.1%

1%

I0*4

I0'

2

I I0

2 10* I0

8 10* I0W

I0'

2 io

14 DC resistance, SI

M5-1-MESTIND-67

111
FIGURE 3

I©- CP.RDC 4/67

?I06

H

9) I,06

H

0)o

= 2

Research Bridge

( 1592 Hz

)

FIGURE 4

y Expected Accuracy of

Calculable Capacitor

Transformer Ratio-Arm Bridge

rResistance Ratio-Arm Bridge

/

^

eOtolOOOHz^^

^

Experimental Bridge

MOOtolOOOHz)

-9

0.0001

ppn

•0.01 ppm

-1

100

ppm

A - for sealed copacltors ( 1kHz ) 1 to 10 kHz

B - for unsealed capacitors ( 1 kHz)

SO

12 10 10 k56 -4

10 !0 10 10

-1%

Capacitance, pF

CALIBRATION OF STANDARD INDUCTORS

CP 8/13/64 _ I I I I

|

I I

|

I | I I I

I

|

I I I l|—| II I|—I I I I |

I I I I

|

I I I I

lOOHZ-i

400HZ

X)-Q—O-O—

Q

— 10kHz

<§) ®—

0.001%

0.01%

I000HZ

•0.1 TOIOkHZ

lulJ m iiI i i i 1 1 iiiiI i i i 1 1 i i i 1 1 i i i 1 1 i i i r

0.1%

icr- 10-2 10" 10 101%

2 INDUCTANCE, H

M5-1-MESTIND-67

112
FIGURE 5

WJH/CP/FRK/FLH - 12/20/66

in

1 1

1

1 1 1 1 1 11 1

1

1 <NBS>

O.OI PPM

c

I0*

7 Comparison of

Std. Cells

0.1

part I0*

6 —

/ •

1

(one

I0*

5 \ Volt Boxes / HV Dividers 10

icertair I0*

4 Potentiometers

\

100

10 1000

io

,!

1 ! 1

i i I l I

If

1

I0"

2 10"' 1 10 10* I0

3 I0

4 I0

5 10

s DC voltage,

FIGURE 6

I I I I I I I 1

I0"

3 I0"

2 10"' I 10 10* I0

3 I0

4 I0

5 I0

8 AC voltage ,V

M5- l-MESTIND-67

113
FLH 3/67

-0.01 ppm

NBS -0.1

-1

+3

10-

DC Resistors

( For use with pot'r

)

AC - DC Transfer Stds.

5 Hz to 50kHz^

NBS Research Transf's Current Comparators

60Hz

~£-

400Hz ^

Current Transformers

-10 ppm

100

r-IOOO ppm

~~T~ K5» 10 1 10

Current,

A

i 0

s 10'

M5-1-MESTIND-67

USCOMM-MBS-DC
ABSOLUTE VALUES OF ELECTRICAL UNITS - I Realization of Electrical Units: Classical Methods

The NBS Absolute Ampere Experiment

P. T. Olsen, V. E. Bower, W. D. Phillips, E. R. Williams,

and G. R. Jones, Jr. (1984) 117

The Realization of the Ampere at NBS

Paul T. Olsen, Marvin E. Cage, William D. Phillips and

Edwin R. Williams (1980) 124

A Proposed Coil System for the Improved Realization of

the Absolute Ampere

P. T. Olsen, W. D. Phillips and E. R. Williams (1980) 128

New NBS Measurement of the Absolute Farad and Ohm

Robert D. Cutkosky (1974) 144

The Fundamental Electrical Standards: Present Status

and Prospects for Improvement

Robert D. Cutkosky (1971) 149

115

The NBS Absolute Ampere Experiment

PAUL T. OLSEN, member, ieee, VINCENT E. BOWER, WILLIAM D. PHILLIPS,

EDWIN R. WILLIAMS, and GEORGE R. JONES, JR.

Abstract-Viz have constructed a current balance with superconducting field coils for the realization of the SI ampere by comparing mechanical to electrical work. The estimated ultimate accuracy of the

realization is 0.1 ppm. We describe and present preliminary results obtained with a room temperature version of the apparatus.

I. Introduction

THE practical electrical units can be maintained by a national standards laboratory, such as NBS, with remarkable

accuracy and precision. The unit of voltage, defined in terms of the ac Josephson effect, is stable in time and routinely maintained to a few parts in 108 [1], In the same way, the

unit of resistance could be defined in terms of the quantum

Hall effect, with a reproducibility and long-term stability of about four parts in l'O

8 [2] . Unfortunately, the relationship between these "as-maintained" units and the more fundamental

SI electrical units (defined only in terms of the units of mass,

length, and time) is not nearly so well established. The SI definition of resistance can be related to the practical defini- tion by calculable capacitor experiments [3] to a few parts in 108

, but the SI definition of voltage is related to the practical or laboratory unit to an accuracy of only a few parts in 106

or worse [4]

Historically, the relationship between the SI and practical units of voltage has been determined by a realization of the SI

Manuscript received August 20, 1984. The authors are with the Electricity Division, National Bureau of Standards, Gaithersburg, MD 20899.

definition of the ampere (A): "the ampere is that current which,

when passed through two parallel, infinitely long wires of negligible cross section 1-m apart, produces a force between them

of 2 X 10"7 N/m of length." The comparison of this SI ampere to the practical ampere, defined by the practical volt and ohm, completes the link between the SI and as-maintained systems of electrical units. There are also various indirect methods for arriving at the

ratio of the SI to as-maintained amperes. Generally, these in- volve combining a number of measurements of fundamental

constants, such as the Faraday, the Avogadro constant, the gyromagnetic ratio of the proton, the Josephson frequencyvoltage ratio and others, some of which involve electrical measurements and some of which involve force measurements. A comparison of such indirect measurements of the SI ampere

with direct realization of the definition indicates large discrep- ancies [4] ; in fact, the value for the ratio of the SI to labora- tory ampere KA

= /I lab/A given in the latest (1973) adjust- ment of the fundamental constants [5] may be in error by as much as 10 parts per million (ppm). The traditional method for realizing the SI definition of

the ampere has been to construct a set of coils of accurately known dimensions, calculate from the dimensions the force which one coil system will exert on the other for a given cur- rent in each, and then to compare the calculated force to the

actual force, as measured using a balance, when a current known in as-maintained units is passed through the coils. In

this case we have

U.S. Government work not protected by U.S. copyright

117
'NBS r/1/2 U,2

C(R)

Making the substitutions / = KAIms and V - KAKn Vws , 0) where Ka = S2N bs/^» we nave

where Ij and 72 are the currents, in laboratory units, in each

of the coils, Fl2 is the force between them, and C(R) is the calculated geometrical factor based on measurements of the dimensions and separation of the coils. There are two key problems in performing this traditional

realization of the ampere. First, the dimensions of a physical

object such as a coil of wire are notoriously difficult to measure to accuracies at the ppm level. The problem arises in large part from difficulties with either mechanical or optical measurements of dimensions: one cannot accurately locate surfaces of objects which are at some level deformable and

reflect light with unknown phase shifts. Furthermore, even

if the mechanical dimensions of a coil were accurately deter- mined, the electrical dimensions would still be uncertain be- cause of the nonuniform distribution of current caused by

strains or impurities in the wire. Second, since accurate geometric measurements of each

wire in the coils are needed, the coils must be wound in a single layer. This limits the magnetic field and thus the force which the coils can produce. Typical ampere balances operate with forces equivalent to only a few grams. Because the bal- ance used to measure this force must carry the large dead

weight of the coil as well as operate in the presence of convective air currents driven by the warm coils, weighing uncertainties are a significant limitation to the achievable accuracy.

In order to avoid the uncertainties of dimensional measure- ment, a new method [6] for realizing the ampere has been

proposed. Consider an ampere balance as two coil systems,

coil 1 (the field coil) being fixed in space and coil 2 (the sus- pended coil) being allowed to move in the field of the first

coil. The force exerted on coil 2 by coil 1 is F12 (z) = /,/2 -dMn (z)ldz (2) where Ix and /2 are the currents in the two coils and Mn is the mutual inductance between the two coils. Only the z component of the force is considered since only vertical forces can be measured (i.e., compared to gravitational forces on known masses) using a balance. If we now consider that coil 2 carries no current, but moves vertically in the field of coil 1, a voltage will be induced across coil 2 in which

V = Il dMl2 /dt. (3) Combining (2) and (3), assuming that Ii is a constant, and

that the functional form of Mn (z) does not change, we have

F{z)

V

h

dz/dt

(4)

where dz/dt is the vertical velocity of the coil under the conditions described for (3). If we integrate (2) and (3) over in- tervals such that the endpoints Zx and Z2 for the integration of (2) correspond to the positions of the suspended coil at the endpoints f, and r 2 for the integration of (3), we have

I F(z) dz

z.

V(t)dt.

• 1 In both (4) and (5), all electrical quantities are in SI units.

(5)

F(z) dz

V{t) dz

1/2

(6)

which replaces (1) for the realization of the ampere. Note

that (5) is simply a statement that mechanical work equals

electrical work.

Several features of (6) are particularly attractive: No coil dimensions are required, and only the displacements of the

coil with respect to an arbitrary reference plane need be measured. This type of measurement is easy to accomplish inter- ferometrically and is not limited by uncertainties in optical phase shifts or in the nature of current distributions. The only

requirement is that the effective geometry for the force measurement be the same as for the voltage measurement. Since no geometric measurements other than the displacement of

the suspended coil are needed, both the field coil and the sus- pended coil can be multilayer coils, or even superconducting

as long as the superconducting current paths do not change

significantly during a measurement. In this way both large forces and large voltages can be produced making accurate measurements of these quantities easier, and the realization of the ampere by (6) will no longer be limited by the same

dimensional uncertainties that restrict realization by (1). In general, there is no fundamental difficulty in achieving

forces and voltages large enough to be measured with an accu- racy of 0.1 ppm or better, and displacements of a few centi- meters are routinely measured to such accuracy. Thus it is realistic to assume that the new ampere realization can achieve an improvement of about two orders of magnitude over the accuracy achieved with present current balances.

II. Apparatus

We have constructed an apparatus for realizing the ampere

according to (6). The mechanical portion of the apparatus is shown in Fig. 1 . The field coils B are rigidly connected to the marble platform H which is attached to a massive limestone

pier J. Two complete versions of this coil system have been

constructed. One, designed for preliminary testing, is made

from copper wires wound on an aluminum form. With a cur- rent of 8 A it produces a radial field of 2.9 X 10"3 T at the location of the suspended coil C. The second version is made

of 20 superconducting coils wound on glass-epoxy forms, and

produces a field of 0.20 T with a current of 10 A. The current

distribution in the two versions of the field coils is nearly identical, so they have similar properties except for the magnitude

of the field. The copper coils are immersed in a temperature

controlled oil bath which removes the nearly 3 kW of heat dis- sipated. The superconducting coils are contained in a liquid He cryostat which consumes 1-2 liters of He per hour. The

design of the coil system is described in more detail in [7]

.

The suspended coil C is made of 4956 turns of copper wire wound on a fiberglass-epoxy form. It has a mean diameter of

118
A

Fig. 1. A simplified scale drawing of the NBS absolute ampere experi- ment. The separate parts are identified, as follows: A. The balance is an aluminum disk or wheel approximately 61 cm

in diameter by about 2.5 cm in thickness. At the hub is a silicon carbide knife the edge of which rests on a boron carbide plane. A

thin brass band strung over the top of the wheel is connected to three pronged "spiders" on each side. The scale pans are in turn attached

to the spiders. B. The magnetic field coil consists of four separate windings. The large upper and lower sections provide the main magnetic field. The axial

fields are in opposition so at the mid-plane of the coil the magnetic

field is in the radial direction. The smaller inner coils (compensation

coils) add some field shaping to the primary radial field such that the

radial field varies inversely with the radius. C. The movable suspended coil or MSC has a mean diameter of 70 cm. It is bifilar-wound in three coplanar sections producing a

coil with a square cross section, 1 cm on a side. The change in force on the MSC upon reversal of the current through it is counter-balanced by an appropriate known standard mass placed on the right-hand

scale pan. An uncalibrated mass on the left-hand scale pan provides

for approximate equilibrium of the balance during the velocity- EMF portion of the measurement. Attached to the MSC coil are the moving arms of the multi-axis laser interferometer which monitors

the vertical displacement for both parts of the experiment (i.e., the velocity-EMF measurement and the force-distance measurement).

D. The fixed suspended coil is wound identically to the MSC and is suspended from the marble platform H. This coil is used to monitor

the unwanted movement of the fixed field-producing coils and sup- ports the fixed arms of the laser interferometer.

E. The right-hand spider, connected to a thin brass strap, wrapped

around the rim of the balance wheel, supports the MSC and the scale pan on which the known standard mass is placed. F and G. The auxiliary magnet coil, F, is a scaled down, room temperature version of the main magnet coil. This magnet, together with

the auxiliary drive coil, G, controls the motion of the balance during

the velocity-EMF portion of the measurement.

H. The massive marble slab is cantilevered on heavy wooden beams

from the top of a substantial limestone pier J. The platform is quite

stable and is as mechanically quiet as any other site at the NBS facility. The balance and all components above the marble platform are housed in a styrofoam insulated wooden cabinet. The suspended

coils are housed in a smaller wooden cabinet suspended from the marble slab by three aluminum tubes. Inside the three tubes are three more tubes of smaller diameter which support the fixed suspended

coil. Inside these three tubes are three other tubes which support

the movable suspended coil from the spider which in turn is attached

to the brass strap around the balance wheel.

70 cm and a resistance of 9 k£2. When a 3.33-mA current is reversed in the suspended coil the force changes by the equivalent of 15 g when the copper field coils are used, and 1000 g when the superconducting field coils are used. When the sus- pended coil is moved vertically at 0.667 mm/s, the generated

voltage is 20 mV for the copper and 1 V for the superconducting coils. The suspended coil fits around the outside of the

oil bath or cryostat containing the field coils. The field coils B, including the additional trim windings

near the midplane of the system, have been computer designed

to optimize simultaneously a number of parameters of the magnetic field at the position of the suspended coil. The force

(or generated voltage) on the suspended coils varies by only 180 ppm for a 6-cm vertical displacement, and the force has extrema at the center and endpoints of the displacement. The

force varies by less than 15 ppm for a 1 -percent change in the diameter of the suspended coil, and is also relatively insensitive to small transverse or angular displacements of the suspended

coil. Because of this insensitivity to changes in the suspended

coil geometry and position, reasonable care will ensure that the geometry during the force measurement implied by (2)

will be similar enough to that during the voltage measurement

implied by (3) that they may be combined to obtain (6). The

force does depend strongly on the geometry of the field coils, so one must rely on the temperature stability of the oil bath

or cryostat, and of the mechanical support structure to maintain this constant. The suspended coil is attached by means of a spider E to the balance A which measures the force and provides the dis- placement of the coil. The balance is in the form of a wheel whose central pivot is a knife edge. A metal band 2.5-cm wide and 5 X 10"3 -cm thick hangs over the rim of the wheel and

supports the spider and suspended coil on one side and an

auxiliary spider and coil on the other side. The use of this wheel and band configuration rather than the more usual beam

and knife edge suspension allows a vertical displacement of

the suspended coil with virtually no horizontal displacement. This is important because the balance can only measure vertical forces (since it compares the magnetic forces to known

gravitational ones). Thus while horizontal forces do not contribute to the measured force, horizontal displacements do

contribute to the generated voltage, so that combining (2) and (3) is no longer valid. In our experiment the symmetry

of the construction ensures that horizontal forces are small, and the use of the wheel and band ensures that horizontal displacements are small. Together, this makes the error from

these causes acceptably small. A miniature version F and G of the coil system B and C

hangs from a spider on the opposite side of the balance. These

auxiliary coils provide the force needed to drive the balance and translate the suspended coil during the voltage measure- ment phase of the ampere realization. With a fixed current in the auxiliary field coil F, a varying current in G applies a vari- able force to the balance. The auxiliary suspended coil G is would in such a way that it links no flux to the suspended

coil C. Thus as the current in G changes, there is no voltage induced into C from this extraneous source. A "fixed suspended coil" D, would identically to the "moving suspended coil" C, is suspended rigidly from the marble slab

119
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

H. The two coils are connected in opposition so that voltages induced by changes in external magnetic fields or by motion

of the field coils B are canceled out. Coil D is also used to define the reference plane for measurement of the displace- ment of coil C. Ideally the reference should be provided by

the coils B, but this is difficult in practice since they are in an oil bath or a cryostat. By measuring both the position and

voltage of coil C with respect to coil D we effectively eliminate any effect from motion of B since its position is electrically transferred to coil D. The entire balance assembly, including the wheel and the suspended coils, is contained in a thermally isolated, double "box within a box" enclosure. The dewar or oil bath containing the field coils is not within the inner enclosure, but

its surface temperature is maintained close to that inside the enclosure. The temperature gradient inside the enclosure

is arranged to minimize convective air currents. Balance operations such as changing weights are accomplished remotely by

hydraulic drives, so that the balance enclosure need never be

entered after the balance is assembled. Damping mechanisms

inside the enclosure reduce swinging and torsion pendulum

motions of the suspended coil. The force measurement phase of the ampere realization is accomplished by alternately removing and replacing a cali- brated mass on a scale pan attached to the spider E, while

servoing the position of the moving suspended coil to a constant displacement from the fixed suspended coil. The servo

is accomplished by varying the current in the suspended coil. The balance is arranged so that removing or replacing the mass

results in a reversal of the current in the suspended coil. For

our preliminary measurements we infer the displacement of

coil C with respect to D from the displacement of the pan

attached to the spider with respect to the marble slab H. Ul- timately the displacement of C compared to D will be measured directly.

Fig. 2 shows the force measurement and servo circuit. A

desk-top computer (DTC) measures the position of the pan

as determined by the laser interferometer system. The DTC

computes the difference (error) from the desired position as well as the time derivative and integral of the error. Numbers

proportional to the error and its integral are summed and

converted to a voltage in the digital-to-analog (D/A) converter. This voltage drives a voltage to current converter (Al) which

supplies current to the suspended coil through a calibrated 300-£2 resistor ^?3, with the current returned to rl. In a similar manner, the derivative of the position error is applied to the auxiliary suspended coil on the other side of the balance. The current in the main suspended coil is measured by comparing the voltage drop it produces in R3 to the voltage drop

in Rl. Rl is a standard l-k£2 resistor supplied by a stable 1-mA current source. The difference between the voltages on

and R 1 is measured by the calibrated detector D, which

in turn is read by the high speed data acquisition system

(HSDAS). The HSDAS also records the position of the sus- pended coils. When the mass is removed or replaced, the servo system reverses the current in the suspended coil, so as to keep

the position of the coil constant. Accordingly, the 1-mA cur- rent in Rl is reversed to keep the difference voltage seen by

the detector small. The 1-mA current source is calibrated by

comparing the voltage on R 1 to the voltage of a standard cell which is ultimately calibrated in terms of the Josephson effect. The current in the auxiliary suspended coil, which is pro- portional to the derivative of the position error, is not measured, because it averages to zero. This separation of the derivative feedback signal reduces the noise in the measurement

of the current in the suspended coil and has much the same

effect as filtering. Fig. 3 shows the circuit for the voltage measurement phase

of the experiment. The voltage induced in the suspended coil

as it moves is compared to the voltage drop in the 20-Sl resis- tor R2, which is driven by the same 1 mA current source used

for the force measurement. The difference (error) voltage is

120
OLSEN etal.: NBS ABSOLUTE AMPERE EXPERIMENT

amplified by detector D and converted to a digital form by

the A/D converter. The DTC reads this error signal, and computes its time derivative, offset, and time integral. Numbers

proportional to each of these are summed and converted to

a voltage in the D/A converter, and to a current in A2. This

current is applied to the auxiliary coil which drives the balance

in such a way as to keep the error voltage near zero. Thus the induced voltage in the suspended coil is held nearly constant by controlling its velocity via the servo loop. One could, of

course, servo the velocity to a constant value and measure the changing voltage, but we prefer to servo the voltage so as to avoid uncertainties from measuring a changing voltage with

a detector having a long time-constant. While the DTC measures the error voltage on the suspended

coil, it also monitors the coil position. When the coil approaches

the end of its range of motion, the 1-mA current source is in- structed to slowly reverse the current in R2, which in turn causes the velocity of the coil to reverse under servo control. During the period between reversals, the HSDAS records measurements of the time and of the position and error voltage of

the coil. About 5000 such measurements are made during the 60 s required for the coil to cover its range of travel. For both the force and voltage measurements the reversals (of current in the suspended coil or of direction of velocity)

serve to cancel the effects of zero offsets in the detector or data acquisition system, and of thermal EMF's. These drop

out when one calculates the change in current or voltage upon

reversal.

III. Experimental Procedure

As of this writing we have only taken data using the copper,

oil-cooled field coils. The procedures for taking data are partially automated and are still being refined. Because of the expense of the liquid He required to run the superconducting

coils, we intend to perfect our procedures using the copper

coils before beginning to use the superconducting ones. At the beginning of a measurement the 1-mA current source

is calibrated against a standard cell. The suspended coil is at the bottom of its travel, held down by the calibrated mass. This defines the zero position of the coil. The standard mass

is lifted from the pan on the suspended coil side of the bal- ance, and the detection and feedback circuit is configured for voltage data as in Fig. 3. The DTC automatically and repeat- edly runs the suspended coil up and down over its travel while

the operator instructs the HSDAS to record time, position, and voltage error for a predetermined distance after each ve- locity reversal occurs. When a sufficient number of traversals are accumulated, re- quiring 5-15 min of real time, the system is reconfigured for force measurement as in Fig. 2. The calibrated mass is lowered onto the pan on the suspended coil side and a counterweight,

half the mass of the calibrated mass, is placed on the opposite

pan. During the changeover the 1-mA current source is recali- brated. Within 3-5 min after the voltage measurement ends, we begin force measurements.

Readings of the position and current of the suspended coil are made for about 60 s, after which the calibrated mass is

lifted slightly off the pan. When the position of the suspended

coil has stabilized, the current in the suspended coil has re- versed under servo control and the reference current has been

reversed manually, readings of coil current and position are resumed. About 100 s of real time separates the start of "mass on" and "mass off" measurements. After a total real time

similar to that spent on voltage measurement, the system is again configured for voltage measurement, with a recalibration of the 1-mA current source during the changeover. The alternation of voltage and force measurements continues for several hours, which constitutes the data for one

day. In principle, the evaluation of A'A using (6) would re- quire force measurements at a number of different positions

z of the suspended coil. In fact, we have taken force measure- ments at only one position, and used the extensive voltage measurements to infer how the force would vary over the range of motion of the coil. Ultimately the validity of this procedure will be confirmed by actual measurement of the

force at different positions. The data are analyzed so as to cancel the first-order effects of drift in the experimental conditions, as well as to cancel most effects of thermal EMF's and zero offsets. For the volt- age measurement data, the time integral fVdt is computed

from the known reference voltage, the measured voltage offset or error voltage, and the times. The range of integration is between two fixed points in space. Tins is done for each set of data constituting a single traversal of the suspended coil in a given direction. The integrals for two "up" traversals are used to calculate an interpolated value for the up integral at the mean time for the down traversal which occurred between

them. The difference between the interpolated up integral and

the down integral gives a value for the time integral of the volt- age which is free to first-order of thermal and zero offsets and drifts. The interpolation and subtraction procedure is continued for all the data in a given group of voltage measurements. Typically, the standard deviation of a set voltage-time

integral differences, taken over 5-15 min, is less than 5 ppm.

This is a measure of the minute to minute scatter in the volt- age phase of the measurement. The expected effect on the value of KA is half of this because of the square root in (6). The force measurements are analyzed in a similar way. The

current is calculated from the known reference voltage, the value of the resistance R3, and the measured error voltage.

All the readings for a set with the suspended coil current in a given direction are averaged and the mean time of measure- ment is computed. Interpolated differences between the opposite current directions are calculated as in the case of

voltage measurements. Differences obtained in this way are

free to first order of drift in the balance point of the wheel, and thermal or zero offsets and drifts. A set of current dif- ferences obtained over a 5-15-min period typically has a stan- dard deviation of less than 5 ppm, representing the minute

to minute variation of the force measurement, which contrib- utes half that uncertainty to KA . The averages of the interpolated force and voltage measure- ments are now combined to yield a value of KA which is free of drift in the coil characteristics such as current in the field

coils, or geometry changes in the coil system. This is done,

121
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1935

for example, by interpolating a value for the voltage integral

at the mean time of a force measurement. Following (6), the interpolated voltage integral difference is multiplied by the average suspended coil current difference on reversal, multiplied by £2NBS /£2, and divided into the product of the cali- brated mass, m, the acceleration of gravity, g, and the distance over which the voltage integrals are calculated. A correction

is made for the expected variation with z in the force mg based on the measured variation of the voltage integral over segments

of the total path traveled by the coils. In this way a value for KA is obtained for every pair of volt- age measurements with a force measurement between and

every pair of force measurements with a voltage measurement

between. These interpolated results are free to first-order of such changes as drifts in the current in the field coils or geo- metric changes in the coils. The standard deviation of a set of such measurements of KA taken on a single day is on the order of 2 ppm. This within-a-day scatter is an important measure of the ability to detect systematic errors: errors more

than several times smaller than this scatter will be difficult to evaluate experimentally in a short period of time. Day to day scatter of the value of KA is of similar importance, and

is thought to be of the order of 3 ppm, although at this stage there is not sufficient data to adequately evaluate this scatter. Day to day variations include the effects of resistor and volt- age reference drift which we have only recently brought under

control by using good quality standard resistors and standard

cells.

IV. Systematic Errors

Before a value for KA may be given, a number of possible systematic errors need to be evaluated. Below, we list some

of the errors which may affect the ampere experiment and

which we intend to evaluate experimentally. One such error is balance motion. Effects from balance motion may arise in two ways. Horizontal motion of the sus- pended coil makes the assumption of vertical forces implicit

in (2) invalid, as has been discussed previously in this paper. We will need to measure the magnitude of any horizontal motion and the size of any associated horizontal forces. Motion

of the suspended coil also exerts mechanical forces on the balance wheel which oscillates with the periods of the swinging and torsion motions of the coil. These motions do exist, and

it must be confirmed that their effect averages to zero, or a correction needs to be applied. There is a relatively large drift in the balance point of the wheel, equivalent to as much as 100 ppm in an hour or so. With such a large drift a first-order drift correction may not

be sufficient. The drift is probably caused by an air temperature change between the upper and lower portions of the bal- ance enclosure, which results in a change in the buoyancy of

the main suspended coil which hangs lower than the auxiliary suspended coil on the other side of the balance. This effect occurs even when the density of the two coils is matched to avoid drifts when the average air density in the balance enclosure changes. The effects of leakages and thermal offsets in the calibration

of the current source used to provide reference voltages need

to be carefully evaluated. The effect of measuring the displacement of the scale pan

rather than of the suspended coil must be evaluated, or the system changed to measure the correct displacement. The effect of nearby magnetic materials remains to be eval- uated. Most magnetic materials are excluded from the labo- ratory, but some, such as the laser, remain. Leakage between windings of the suspended coil can cause an error if this leakage depends on the voltage applied, since

this would result in a different effective geometry for the force and voltage phases of the measurement. When the superconducting windings are used, it will be nec- essary to determine the effect of the susceptibility of the superconducting wire. Any residual effects due to coupling between the auxiliary

drive coils and the main suspended coil must be measured.

V. Expected Results

With the present configuration, using the copper field coils, we have achieved a statistical uncertainty on the order of 3 ppm for about an hour's worth of real time spent accumulating data, with day to day variations expected to be 3 ppm.

With this apparatus it should be possible to evaluate systematics at the 2-ppm level or better, yielding a value of KA accurate to that level. When the superconducting coils are used,

the voltage and force will increase by about a factor of 50. This should reduce the statistical error due to fluctuations in these quantities by that same factor. We believe that most

of the scatter which we experience falls into this category. Among the errors which will not be reduced when we use

the superconducting system are those related to the distance measurement and those related to short-term fluctuations in current supplies. We do not foresee any difficulty in having

the fluctuations in the current supplies below the 0.1 ppm

level, and our experience with the interferometric distance measurements indicates that measurements better than 0.1 ppm over 6 cm should not be a problem. Thus we expect

that in the superconducting version of our ampere experiment we should achieve a statistical precision better than 0.1 ppm.

Until we have examined all of the sources of systematic

error, it will be impossible to predict with confidence the accuracy limit we will ultimately achieve. Nevertheless, we

do not see any indication that errors significantly greater than

0.1 ppm will limit the final result. Furthermore, the projected

statistical precision gives us confidence that we will be able to evaluate the systematic errors at the sub-O.l-ppm level. In conclusion, we expect that the present experiment with copper field coils will yield a value of KA within an uncertainty by one to a few ppm, while we expect to achieve a final accuracy of 0. 1 ppm when the superconducting field coils are used. It is even possible that refinements in experimental

techniques will allow us to reach uncertainties in the 0.01 -ppm

range, which opens the interesting possibility of using a current balance and the as-maintained electrical units to monitor the

stability of the unit of mass.

122
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985 181

Acknowledgment

The authors would like to thank J. A. Hammond and M. Zumberge for measuring the acceleration of gravity, g, at the

site of our ampere balance; R. S. Davis for calibrating the mass

used; R. F. Dziuba for calibrating the resistors; J. E. Sims for calibrating the standard cells; and G Barrett for assistance in taking the data.

References

[1] B. F. Field, T. F. Finnegan, and J. Toots, "Volt maintenance at NBS via 2e/h: A new definition of the NBS volt," Metrologia,

vol. 9, p. 155. 1973.

[2] M. E. Cage, R. F. Dziuba, and B. F. Field, "A test of the quantum

Hall effect as a resistance standard," IEEE Trans. Instrum. Meas.

pp. 301-303, this issue. [3] R. D. Cutkosky, "New NBS measurements of the absolute farad and ohm "IEEE Trans. Instrum. Meas., vol. IM-23, p. 305, 1974.

[4] B. N. Taylor, "Is the present realization of the absolute ampere in error?" Metrologia, vol. 12, p. 81, 1976.

[5] B. N. Taylor, W. H. Parker, and D. N. Langenberg, Rev. Mod.

Phys., vol. 41, p. 375, 1969; and E. R. Cohen and B. N. Taylor,

/. Phys. Chem. Ref. Data, vol. 2, p. 663, 1973.

[6] B. P. Kibble, "A measurement of the gyromagnetic ratio of the proton by the strong field method," Atomic Masses and Fundamental Constants, vol. 5, J. H. Sanders and A. H. Wapstra, Eds. New York: Plenum, 1976, p. 545.

[7] W. Y. Chen, J. R. Purcell, P. T. Olsen, W. D. Phillips, and E. R.

Williams, "Design and construction of a superconducting magnet

system for the absolute ampere experiment," in Advances in Cry- ogenic Engineering, vol. 27, R. W. Fast, Ed. New York: Plenum,

1982, p. 97.

123
IfcEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 1M-29, NO. 4, DECEMBER 1980 The Realization of the Ampere at NBS

PAUL T. OLSEN, member, IEEE, MARVIN E. CAGE, WILLIAM D. PHILLIPS, AND

EDWIN R. WILLIAMS

Abstract

We present a method for the realization of the ampere

based on Faraday's induction law and using a modification of the

classic Pellat balance. A preliminary apparatus has been constructed and initial measurements have been obtained. This balance is also compared with a balance similar to one proposed earlier.

Introduction

THROUGHOUT the history of the National Bureau of Standards (NBS), the direct measurement of the NBS

ampere in terms of the fundamental units of mass, length, and

time has been periodically performed. The last such measurement occurred in 1968 [1], In the past, these measure- ments have been essential to establish an accurately known

national voltage reference. At present, a more accurate ampere

determination is essential to help resolve discrepancies that

exist among the measured values of some of the fundamental

physical constants [2]. The NBS effort to carry out a sub-parts per million (ppm)

ampere determination is now in progress: a new coil system has been designed and construction will soon begin; a multi-axis

laser interferometer system has been obtained and tested; a data acquisition system designed for remote operation and

high-speed data rates has been installed; and a preliminary ampere balance in the form of a Pellat "electrodynamometer"

has been assembled and is now in operation [3]. This report

describes the preliminary ampere balance and briefly discusses the resolution of this instrument.

Theory of the Measurement

Following a proposal by Kibble [4],' the need for measuring

the dimensions of a physical object has been replaced by the need to effectively measure the change in mutual inductance between two coils while they move relatively to each other. A

Pellat-type balance is an especially good instrument with which

to study and develop appropriate measurement techniques. It has a current-carrying coil which is permitted to rotate in an

externally applied magnetic field (usually established by means

of a current in a solenoid or Helmholtz pair) and which expeManuscript received July 21, 1980 This project was supported by the National Bureau of Standards. The authors are with Electrical Measurements and Standards Division, Center for Absolute Physical Quantities, National Bureau of Standards, Code

521, Washington, DC 20234.

1 In addition to Kibble's paper two other ideas are also reflected in the present Pellat experiment. In 1972, D. S. Sullivan suggested measuring the

integral of / times t and comparing it to the mechanical work done on a su- perconducting motor [7]. Also, C. H. Page in 1971 suggested to one of the authors, E R. Williams, that the derivative of the mutual inductance with

respect to the balance pan displacement is a constant for the Pellat bal- ance.

riences a torque tending to align its axis with that of the magnetic field. The torque on the rotatable coil is given by

7(0) - Iih(dM12/d6) where I\ and I 2 are the currents in the rotatable and external

circuits, respectively, M\ 2 is the mutual inductance, and 0 is the rotation angle. The integral of this torque in moving the coil between two

angles (initial and final) is then W = T(6)dd = /,/2 dM

- I\h[Mn(9f) ~ Mi2(6d]. (2)

If the coil is now allowed to rotate with its circuit open, the magnitude of the EM F generated is

€(0 = I 2 (dM/dt). (3)

Integrating this EMFover time (from time t\ to time t 2 ) one

obtains

C'

2 e(t)dt = h C'dM = I 2 [M n(t 2 ) - A/ 12 (/.)]. (4)

/, =

If /| and ?2 are the times when the coil is at angles 0,- and 0/, respectively, of (2) we may divide (2) by (4) to obtain

T(6)de J

§\(d)dt. (5)

Since the numerator of (5) is just the work done by the

electrical torque in rotating the coil between 0, and 6j, it can

be replaced by any equivalent expression. In particular, this work is equal to the work done by a vertical external force applied at the end of a balance beam when that external force

just balances the magnetic force on the movable coil. This force can be measured experimentally by weighing. Expressed this way, (5) becomes

h= 1j*

f Fz {z)dz /§'f

e(t)dt (6)

where z, and zj are the vertical positions of the end of the balance arm at angles 0, and 0/, respectively. Equation (6) can be written explicitly in terms of NBS

electrical units so that we finally obtain KA 3 (Asbs/A) = ^f

2/ F(z)dz

UNBs(^NBs/fi) I f NBS( t)dt

1/2 (7)

124
OLSEN el a!.: THE AMPERE at nbs

Fig. I . The general appearance of the Pellat-type ampere balance with the

rotating coil B, the central knife edge C, and the standard mass E. Added

to the Pellat balance for the present measurement are: the torque coils A

and the corner cube for the laser interferometer system D. An additional torque coil (not shown) is placed inside the rota table coil. The three torque

coils are electrically connected in series. The third coil prevents the torque

coils from inducing an EMF into the rotatable coil.

Fig. 2. The voltage of a reference source is compared to the EMF of the

rotatable coil. This voltage difference is amplified and drives a current into the torque coils. This current provides the torque required to maintain the angular velocity of the rotatable coil such that the EMF produced in the

rotatable coil is constant and equal to the reference voltage.

where F(z) is the product of the acceleration due to gravity and

the mass on the pan at the position z. /nbs is the magnitude

of the current in the movable coil expressed in terms of the NBS unit of current; QNBs/^ is the ratio of the NBS unit of

resistance to the absolute or SI unit of resistance as measured

by the calculable capacitor [5]; and (nbs is the magnitude of

the induced voltage expressed in terms of the NBS unit of

voltage. The 5\{F(z)dz term is computed from many static weighings made along the path from z, to z/ with a current /nbs in the rotating balance coil. The / 'rffNBs(')^ term is measured

in a separate experiment when the coil is moved from position

z, to Zf in a time // — /, at a nearly constant velocity and the current /nbs is not flowing in the rotating coil. The geometry and the current of the stationary coil must remain unchanged

during all these measurements or an appropriate correction must be made.

Measurement of the Voltage Integral

The evaluation of the integral $'t ]t(t)dt establishes all of

the geometrical specifications of the Pellat balance. The time

l\ begins when the balance beam is tilted to the 0; position and

ends at time t 2 when the balance beam has rotated to 0/. In reality, neither 0, nor 6/ is measured, but rather the

vertical displacement of the primary scale pan. The apparatus

is shown in Fig. 1

In actual operation, the induced voltage is held constant via a servosystem, and the time intervals required to travel pre- determined distances are recorded over the path length. There

is less of a problem in measuring the induced voltage with a detector having a long response time if the measured voltage

is held nearly constant. 1

Fig. 3. A mirror mounted on the axis of rotation of the rotating coil reflects a laser beam onto a split photovoltaic cell. An unbalance in the illumination produces a dc voltage across the photo-cell The amplified signal produces

a current in the torque coils which rotates the balance beam until the output

voltage of the photo-cell is zero. This servosystem controls the balance point

to a few microradians.

Fig. 2 shows a simplified schematic of the servo system

which maintains the constant EMF. The balance arm is driven

to rotate about the central knife edge so that the EM F gener- ated in the rotatable coil equals that of the reference voltage. When the rotation nears either end, an optical sensor triggers a reversing switch which inverts the polarity of the reference

voltage causing the balance to reverse its direction of rotation. The laser interferometer is used to trigger a clock reading when

a predetermined number of fringes has been counted. After an initial settling time, the EMF integral can be calculated from the sum of all voltage measurements between any two

time measurements taken between reversal points. Each time a specified number of interferometer fringes is counted, the data acquisition system reads the accumulated

time of a 1-MHz clock and the difference between the refer- ence voltage and the EMF generated in the rotating coil. This

correction voltage, typically less than I j/V with noise of a few

tenths jiV, is also used to drive the servo system. The reference

voltage, about 1 mV and stable to a few ppm, is also monitored

at frequent intervals. The measured reference and correction

voltages are added to give the coil voltage. Data are taken with

the balance swinging in both directions to cancel the effects of voltage offsets and thermal EMF's. Integration of the coil voltage over time is done numerically, and the results for nu- merous pairs of up and down swings are averaged.

Measurement of the Force Integral

Fig. 3 shows the abbreviated schematic of the weighing

servosystem. A mirror located on the rotatable coil reflects a

laser beam onto a split photovoltaic cell. The analog servo- system fixes the balance beam in such a position that the laser beam equally illuminates each half of the photovoltaic cell. This servo system fixes the angle of the balance beam to within

several microradians. A mass of 1 500 mg is added to the scale pan on which the corner cube is mounted. The current in the rotatable coil is adjusted to nearly counterbalance the torque of the standard mass at the end of the balance beam. The servo system provides the additional torque with current in the torque coils. The

torque coils are calibrated frequently by noting the change in current in them for a known change in current in the rotatable

coil. The vertical position of the scale pan is read from the in- terferometer. 5
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-29, NO. 4, DECEMBER 1980

COMPENSATION

COIL

MAIN FIELD COIL

NOT TO SCALE

20.90 cm

J

MAIN COILS

169 turns In 112 layers I 20 mil wire COMP. COILS

32 turns In 21 layers 20 mil wire SUSPENDED COIL

70 cm fills. Ic/IM = .MM407S; IMs 50.94285 amps AFc = 176 ppm 2 AZo= 5.4 cm

(a)

-E

COMPENSATION

COIL

COMPENSATION

COIL

MAIN FIELD COIL

20.44 cm

MAIN COILS

676 turns In 26 layers 20 mil wire COMP COILS

32 turns In 21 layera 20 mil wire SUSPENDED COIL

69.70 cm din Ic/IM .64438562 IM=56.63029 amps AFc =176 ppm 2AZoc5.4cm

(b)

Fig. 4. (a) and (b) The design criteria are given for two possible current balances. The magnetic field producing coils and compensation coils are su- perconducting. The suspended coil carrying 10 ampere-turns produces a vertical force of 1 kg. The force on the suspended coil is to first order independent

of its diameter and of the vertical displacement about the midplane of the field coils. A change in the diameter of the suspended coil by I cm changes

the vertical force by less than 20 ppm. Vertical displacements (z 0 ) of 2.7 cm above or below the midplane produce a change AFC in the vertical force of 1 78 ppm, with currents /« in the field coils and lc in the compensating coils as noted. Both superconducting coil systems shown produce the identically same force on the suspended coil. This permits the manufacturer to select the design which he feels would give the best performance from cryogenic

considerations. 2

: The system described here is similar to that of (6] except that it is simpler to construct and requires a standard type of dewar. 126
OLSEN el al.: THE AMPERE AT N BS

A current reversal in the rotatable coil occurs simultaneously with the removal of the standard mass on the scale pan. Each

set of weighings at a position z constitutes one value of the integrand of fl{F(z)dz. Weighings are then repeated over the range z, to z/ and the results are numerically integrated.

The statistical uncertainty of a single set of weighings at one

position which requires 5 min, including calibration of the servo system, is less than 30 ppm. Data taken over 45 min, including

reversals of the current, are scattered from similar sets by less than 1 0 ppm. We expect a substantial improvement when some

care is taken in the mounting and mechanical isolation of the balance.

Measurement Results

Measurements with the Pellat balance to date have con- centrated on the development of control equipment and measurement techniques. Therefore, we have not evaluated in detail the effect of systematic uncertainties. The scatter be- tween different sets of measurements of voltage and force in- tegrals implies a statistical uncertainty of about 10 ppm in KA.

Discussion

Previously, it was stated that the Pellat balance is a good

instrument to evaluate the recently proposed method [4] of measuring KA The most attractive feature is that there is only one degree of freedom: rotation. The rotatable coil of the Pellat balance, in a uniform magnetic field, has a constant voltage induced in it for a constant vertical velocity of the scale pan.

Similarly, a constant current in the rotatable coil will support

in equilibrium a single mass on the scale pan at any vertical position of the scale pan. Our Pellat balance is a modification of an existing balance. The balance was supported on wooden piers resting on a ce- ment floor. The 1-mT magnetic field was supplied by Helmholtz coils wound on wooden forms. Thus major improvements

are clearly possible. However, a difficulty with the Pellat balance is its sensitivity to changes in the dimensions of the

rotatable coil. Our rotatable coil has 43 000 turns of 0.2-mm

diameter copper wire. Thermal expansion of the copper changes the coil's area by about 40 ppm/°C, and the torque produced varies as the area. The height of the coil also changes

by about 20 ppm/°C, thus affecting the balance sensitivity. Even more critical is the shift of mass on the balance beam as the rotatable coil diameter changes dimensions. The coil is not anchored uniformly on the balance beam so the coil "creeps" along it. Considering that the mass of the rotatable coil is ap- proximately 2 kg and has a diameter of about 1 0 cm, the bal- ance moment could change by as much as 0.4 gm-cm/°C.

Our experience has convinced us that the Pellat balance,

properly supported and with a stronger external magnetic field, could be used to measure KA to the ppm level. Control of the rotatable coil temperature, by always maintaining constant power in the coil, would greatly reduce the effects described

above.

Future Plans

The eventual NBS high accuracy (parts in 10 7

) ampere

determination will use an instrument which more closely re- sembles the classical current balance (see Fig. 4 (a) and (b))

[6]. In this system, the magnetic field coils will be superconducting, while the force coil, suspended from the arm of a balance, will be at room temperature. Because of the high

current densities in the superconducting field coils, both the EM F generated in the moving coil and the vertical force to be measured will increase by about 3 orders of magnitude over

those in the preliminary Pellat balance described above. These

will result in a thousandfold improvement in the voltage and

force measurements. The velocity measurements will be made

in a manner similar to that described for the Pellat balance. Contrary to all previous ampere balances, however, the

vertical force on the suspended coil will be, to first order, in- dependent of its diameter and nearly independent of vertical position for a substantial displacement about the midplanc of

the field coils, i.e., for 5- to 6-cm total displacement.

The swing of the balance beam will cause lateral motion of

the suspended coil as it travels vertically. A balance now under

consideration, however, will servo the vertical path of the suspended coil to eliminate lateral motions by providing the

flat of the central knife with the ability to move in a horizontal

plane. To eliminate the problem of the motion of the cryogenic field

coils, a coil similar to the suspended coil will be prepositioned and fixed around the outside of the dewar. This coil, which

defines a reference plane, will have an EMF generated in it should the field coils move. All voltage and displacement measurements of the suspended coil will be made with respect

to this voltage and this position reference plane.

Acknowledgment

The authors would like to thank J. A. Hammond for measuring the acceleration due to gravity g at our experimental

site. This accurate value will be utilized in our forthcoming

experiment.

References

[ 1 ] R. L. Driscoll and P. T. Olsen, report lo Comite Consultatif d'Eleclricite, Comitc International des Poids et Mcsurcs, I 2th Session, 1968.

[2] B. N. Taylor, "Is the present realization of the absolute ampere in error?" Metrologia. vol. 12, p. XI. 1976.

(3) R. L. Driscoll, "Measurement of current with a Pcllat-typc clectrodyn- amomclcr," J. Res. Nat. Bur. Sid., vol. 60, p. 287, 1958.

[4] B. P. Kibble, "A measurement of the gyromagnctic ratio of the proton

by the strong field method," Atomic masses and Fundamental Constants.

vol. 5, J. H. Sanders and A. H. Wapstra Eds. New York: Plenum, 1976,

p. 545.

[5] R. D. Cutkosky, "New NBS measurements of the absolute Farad and Ohm." IEEE Trans. Instrum. Meas., vol. 1M-23, p. 305, 1974.

[6] P. T. Olsen, W. D. Phillips, and E. R. Williams, "A proposed coil system

for the improved realization of the absolute ampere," J Res. Nat. Bur

Std.. vol. 85, p. 4, 1980.

[7] P. B. Sullivan and N. V. Frederick, "Can superconductivity contribute

to the determination of the absolute ampere?" IEEE Trans. Magnetics.

vol. MAG-13, p. 396, 1977.

127
JOURNAL OF RESEARCH of the National Bureau of Standards Volume 85, No. 4, July-August 1980 A Proposed Coil System for the Improved Realization

of the Absolute Ampere

P. T. Olsen,*W. D. Phillips* and E. R. Williams*

National Bureau of Standards, Washington, D.C. 20234

April 1, 1980

In order to resolve the discrepancies which presently exist between the directly measured values of the absolute or SI ampere and the calculated values obtained indirectly from other fundamental physical constant

determinations, one must design an absolute ampere experiment which will produce a result with an uncertainty

of one half part per million or less. A new approach recently proposed by Kibble promises such sub-ppm accuracy. Presented here is the design and evaluation of a coil system which will fulfill the requirements of this new

approach.

Key words: Absolute ampere; current balance; magnetic force; radial magnetic field; superconducting coils.

1. Introduction

One of the first experiments conducted at the National

Bureau of Standards (NBS) was that of determining the as maintained unit of electrical current in terms of the absolute ampere. To date three different methods have been

employed: the classical current balance, the ratio-of-radii method, and the electrodynamometer method.

1.1. The classical current balance

A pair of equal diameter coaxial coils with their axes vertical and electrically connected so that their axial magnetic

fields are in opposition, produce a purely radial magnetic

field at the mid-plane. A third coil (also with its axis vertical)

is suspended from the arm of a precision balance in this radial magnetic field. The interaction between the magnetic

field produced by the coil pair and the current in the sus- pended coil produces a vertical force on the suspended coil that is measured with the balance. The expected force is calculated from the measured physical dimensions of the

coil system The difference in the actual force from that calculated is an indication of the difference between the

laboratory, or as-maintained unit of current, and the absolute ampere as defined in the Systeme International

d'UnitesfSrHl].1

"Center for Absolute Physic*! Quantities National Measurement Laboratory.

' Figures in brackets indicate literature references at the end of this paper.

1 .2. The ratio-of-radii method

The ratio-of-radii technique is in general identical to the

classical current balance except that the coils do not have to be measured quite so carefully. Only the ratio of the radii of

the coils producing the radial field to that of the suspended

coil needs to be measured. This ratio is determined by

measuring the ratio of currents in the field and suspended

coils needed to produce a null field at the center. The criti- cal dimensional measurement here is of the centrally locat- ed probe used as the zero field sensor since the finite size of

this object significantly influences the accuracy of the ratio

[2].

1 .3. The electrodynamometer method

The Pellat balance or electrodynamometer consists of a small coil mounted on a balance beam. The coil axis is positioned orthogonally to a uniform magnetic field at the center of a precision solenoid. In this technique, the torque on

the small coil is measured. The torque constant of the bal- ance is calculated from the dimensional measurements of

the coils and balance beam [3,4]. Common to all three methods is the small mass used as a

restoring force on the scale pan of the balance, typically one

to two grams. The mass of the system on the balance beam

is approximately one kilogram, so a weighing of 1 part in 109 is necessary in order to just achieve an experimental uncertainty of 1 part in 106 or 1 ppm(part per million). Since a

128
weighing of 1 part in 109 is not readily achievable, the pre- vious experiments were already limited to several ppm uncertainty. This restriction, combined with the tedious meas- urements for determining the actual size of some physical

object, limited the accuracy of all previous experiments to a value considerably in excess of 1 ppm.

In addition to making direct current balance measure- ments of KA , the ratio of the as-maintained ampere to the absolute or SI ampere, it is possible to compute a value of KA through the route of other measured fundamental constants. At the present time, these computed values of KA ex- hibit much less uncertainty than the several direct meas- urements of KA . Yet these derived values of KA generally

disagree among themselves and they in turn disagree with

the direct current balance determinations oiKA [5]. In light of the recent advances in technological tools such

as the computer, laser interferometry, and cryogenic tech- niques, it appears that a conventional current balance ex- periment could be conducted to yield a result forA^, with an

uncertainty of better than 1 ppm. Examinations of past ex- periments indicates that the chances of success would greatly improve if: (a) the electromagnetic force to be measured

was increased by two or three orders of magnitude; and (b) the need for super-accurate dimensional measurements of a physical body of inconvenient geometrical shape were less- ened or eliminated.

B. P. Kibble of the National Physical Laboratory (NPL),

has proposed a new method of conducting an absolute ampere experiment which appears to have the capability of measuring KA to an uncertainty of significantly less than 1.0 ppm [6,7]. Kibble's approach eliminates the need to measure the physical size of coils and allows a three order of magnitude increase in the electromagnetic force. This paper follows Kibble's general approach and presents a unique coil geometry by which the absolute ampere should

be measurable to an accuracy approaching 0.1 ppm.

2. Theory

Although the new method proposed by Kibble has been

published, for the sake of clarity the idea is briefly devel- oped here. A length of wire, I, traversing a uniform magnetic field, B, at a constant velocity, v, has a voltage potential differ- ence £ induced at its ends according to

e = t

• (B x v) = v • (f x B). (1)

When the wire is stationary and a steady current, /, flows within it, a force is exerted on the wire given by

F = 11 x B. (2)

It is sufficient for our purposes to examine the situation

where f, B, and v are orthogonal. Both I and B are the same in eqs(l) and (2), so eq(2) can be written as

F = eI/v. (3)

Applying the above equations to the classical current bal- ance, one can imagine a coil of N turns and radius a„ in a uniform radial magnetic field, Br

. The coil is positioned in the center of and co-planar with the radial magnetic field. The term I can be identified with the coil circumference

times the number of turns on the coiL Imagine now the coil traversing the radial field in an axial direction at a given velocity, vz

. An emf £ is developed across the coil terminals

and is measured in terms of a laboratory as-maintained unit

of voltage. In analogy with eq. (1), its magnitude is given by

£ = tjtU VsbsI V) = N2na0 BrVl (4)

Here, VNBS is the NBS as-maintained unit of voltage, V the absolute or SI volt, and zNBS the induced emf measured in terms of VNBS. Assuming in eq (4) that the velocity v, is measured in terms of the meter and the second, then the product of the number of turns N times the circumference 2na0 times the magnitude of the magnetic field Br is "calibrated" and

given by

N2na0Br = tNB^VNBS/V)/vI . (5)

Now imagine the same coil being suspended in the same

radial magnetic field from a balance beam and carrying a current / measured in terms of the NBS as maintained unit of current, ANBS. The resulting force is then determined by

use of the balance, a calibrated mass m, and knowledge of

the local value of the gravitational acceleration, g. In analo- gy with eq (3) one obtains

F= mg= N2na0 Br I = N2na0Br INBs(ANBs/A). (6)

Using eq (5) we obtain

mgvz = zNBSINB^ V

„

BS/ VXANBS /A). (7)

Noting that VNBS = ANBSQNBs, where QNBS is the NBS as- maintained unit of resistance, yields finally

KA = [Asbs/A) = [ ]" 2 (8)

The ratio of the NBS ohm to the absolute ohm, QNBS/Q,

may be obtained using the calculable cross capacitor to an

accuracy of a few parts in 108 [8]. The quantity g may be ob- tained to the same accuracy by using an absolute gravime129
ter. The remaining terms in the square brackets are meas- ured during the course of the experiment. It should also be

noted that eq(8) can be shown to hold when the force, velocity, and induced emf are replaced by their average values [See Appendix A] [6,7].

3. Design

The classical current balance configuration is inherently

a good design. This configuration contains a high degree of symmetry as well as being a relatively simple geometry to construct. The current balance configuration would lend it- self very nicely to the Kibble approach if it were possible to compensate the radial magnetic field component such that:

(i) the force on the suspended coil is relatively independent

of its axial location (dF/dz ~ 0); and (ii) the force on the suspended coil is relatively independent of its own diameter

2aJidF/da0 ~ 0). The reason for condition (i) is that in the first part of the experiment, the suspended coil is made to traverse the

radial field axially at a constant velocity. Unless dF/dz ~ 0, the generated emf will be difficult to ascertain at any particular instant of time because of the response time of the suspended coil and of the detector system. This situation leads to the desirability of the condition dF/dz = 0, particularly at the beginning and end of its travel. Condition (ii) is desirable because when the velocity is measured, the sus- pended coil will not have a current flowing through it, but

there will be a current present when measuring the force. The heat generated by this current could change the diameter of the suspended coil by as much as 25 ppm/°C. While

a heat source could be used to keep the power dissipation in the suspended coil constant, it would be desirable to mini- mize this systematic error by careful design

With these two fundamental conditions in mind the analytical solution can begin.

The analysis of the classical current balance starts with

the following equation which gives the force acting on one

current loop by another current loop (see fig. 1) [9].

hhzk 2 — k

2 F= [( )E-2K\n. (9)

Here, K and E are the complete elliptic integrals of the first and second kind, respectively. The modulus k is

4oia2 k = [ Y'\ (10) z

2 + (a, + a2 )

2

After some consideration, one soon realizes that the primary

magnetic field producing coils of the classical current ball

J I,

I!

FIGURE 1. Two coaxial loops are separated by a distance z.

Current loop 1 has a radius of a, and a current of /,: Current loop 2 has a radius of a> and a current of /2.

ance cannot simultaneously meet conditions (i) and (ii) stated above. It will thus be necessary to add "compensation coils" in a configuration such as shown in figure 2. Unfortunately, applying the force equation (9) to such a system

and attempting to solve for a configuration satisfying conditions (i) and (ii) is quite difficult Thus it was decided that a computer would be used to examine many of the possible

configurations. After some preliminary calculations the following possibilities were examined, assuming a fixed a2 :

0<a, <a2 0 < Zi to greater than z2 0.2a2 <z2 < 2a2 -10<(/1 //2)< 10

The shaded region of figure 2 was the volume studied to see if solutions consistent with conditions (i) and (ii) existed. The suspended coil of diameter 2a0 would be located some- where in this region.

The immediate results indicated, as would be expected,

that there is not a unique solution but that many solutions

exist. What follows is a description of the solution considered to be the most appropriate for the task intended. For ease of

discussion, 2a2 , the diameter of the outer fixed coil, was

chosen to be 100 cm but note that once the relative values

130
FIGURE 2. A four coil system consists oftwo cod pairs.

The cod pairs are coaxial and concentric The magnetic fields in the axial direction produced by the upper coils in each pair are opposed by the axial direction magnebc fields produced by the lower coils in each pair. The resultant magnetic field in the shaded region is almost entirely directed in the radial

direction. The computer searched the shaded volume for a region where the magnetic field behavior was such that when a current loop, coaxial with the coil

pairs, was suspended in the region, the current loop would have a constant vertical force exerted on it regardless of its diameter or vertical position.

for all coil diameters as well as the separation of the fixed

coils are determined, the entire apparatus may be scaled up

or down without affecting the vertical force exerted on the suspended coil, provided that both the number of windings

and the current in the windings remain constant. On the

scale chosen, the diameter of the suspended coil at which

favorable solutions were found is around 78 cm.

Figure 3 shows the variation in force on the suspended

coil as a function of its vertical position for various separations of the coil pairs. All solutions show a similar behavior.

The z axis is the vertical displacement of the suspended coil about the mid-plane (z = 0) of the fixed coil assembly. The

change in force on the suspended coil relative to that at the mid-plane is designated AF. The positive position at which

LF/dz = 0 is defined to be z0 . Thus, the dashed line is the locus of the points z0. The change in force on the suspended

coil at z0 relative to the force at the mid-plane is designated

AF0. The AF0 and z0 values are determined by separation of

the two pairs of fixed coils and can be adjusted by changing

z, and z 2 , the separation of the fixed coil pairs. The curves a through e show the AF as a function of z. From one curve to the next (a to e), z, and z 2 , the separation of the fixed coils, have been increased in the manner indicated. From eq (3), we can see that the change in emf generated for a constant

velocity is proportional to AF. It is desirable to have AF and

Ac as small as possible for accurate measurements of Fave and At the same time, it is desirable to have large z0,

ou

ata.

V)

C/5o

AF = (F(z)-F(o))/F(o) inppm

FIGURE 3. The change in vertical force on the suspended coil is a function of its vertical displacement about the midrplane of the fixed coil system.

The object of the fixed coil design is to allow a maximum suspended coil displacement for the

least amount of total force change. The dotted line connects the locus of points at which A/7 dz = 0. While A/*o is the force change in ppm at the point where bF/dz = 0 referenced to the mid-plane

force. The A/* as a function of z was calculated for a coil system where a, — 1 1 cm, a2 = 50 cm, /,//a =

5.5438, Oo (the radius of the suspended coil) = 39 cm.

Curve Af„(ppm) ii(cm) z, (cm)

a 10 87.6836 51.5370

b 20 87.8432 51.6166

c 30 87.9702 51.6796 d 40 88.0794 51.7336

e 50 88.1768 51.7816

since the total coil displacement 2z0 must be measured ac- curately. Therefore we must compromise between the desire

for a large z0 and a small AF0. Figure 4a shows the solutions for the separation of the

fixed coils. The abscissa is the diameter of the inner fixed

coil 2a,. The ordinate on the left side is the separation of the outer coils z 2 and the ordinate on the right side is the separate of the inner fixed coils z,. For 2a, greater than 24.0 cm, there is a gradual deterioration of the features desirable

131
1 1 1 1 r Separation of outer fixed colls Z 2 ,AF = 50 ppm

,AF0 =25 ppm

'AF.= 10 ppm - 54

Separation of inner

fixed coils Z1 Afp= 50 ppm

F0 =25 ppm

AF0=10ppm

86

16

cm

18 20 22 24

Diameter of inner fixed coils 2ai cm

FIGURE 4a. For a fixed 2aj, the diameter of the outer fixed coils, the separation of the fixed inner and outer coils z, and z2 can be plotted as a

function oj\2a„ the diameter ofthe innerfixed coils. For a 2a, greater than about 25 cm, li, the separation of the outer fixed coils, increases very

rapidly.

for a working current balance. Specifically, the force becomes increasingly sensitive to changes in the diameter

of the suspended coil. Three curves are shown for each coil separation corresponding to a AF0 of 10, 25 and 50 ppm (as

labeled). The next figure (fig. 4b) shows the working length

or total usable distance 2z0 that the suspended coil can

travel. The abscissa is 2a,, the diameter of the inner fixed

coils. The total usable length is shown for a AF0 = 10, 25,

and 50 ppm. The usable length decreases slightly for an in- crease in 2a, but the change is really insignificant For a further increase in AF0, the total usable length of travel of

the suspended coil does not increase significantly.

2z, cm

AF = 50 ppm

AF0 =25ppm

AF.= 10 ppm

18 IS 20 22

2ai Cm

Figure 4b. The total measurable length 2io as afunction ofAFo> is plot- ted against 2a„ the diameter ofthe innerfixed coils. While there is a decrease in 2z0 for an increase in 2a,, the decrease is not significant

For ease in operation, it would be desirable to have the same current passing through all of the fixed coils. It has

been determined, though, that for a specific force to be ex- erted on the suspended coil for a given current in the fixed

coils, that the number of turns of wire on each pair of fixed

coils is a function of the diameter (2a,) of the inner fixed

coils. Figure 4c gives the required turns ratio (Nx /N2 ) of inner fixed coils to outer fixed coils as well as the number of turns required on the outside fixed coils N2 as a function of

2a,. The abscissa is 2a,. The left ordinate is the turns ratio while the right ordinate is the number of turns on each

outer fixed coil.

N1/N2

3100

N2

3000

2900

1(S '16 m 22 24 26 Diameter of inner fixed coils 2a, cm

FIGURE 4c. The number of turns of wire required on each of the fixed

coils are plotted with respect to 2au the diameter ofthe innerfixed coils. The current in the wire as well as the vertical force on the suspended coil are constant Clearly the most advantageous design would call for an inner coil diameter of about 24 cm For this choice the

total wire necessary for both the inner and outer fixed coil is the least

Figures 4 a, b, and c contain the necessary information

required to build the field coils of a current balance. It was

previously stated that a deterioration takes place for 2a,, in- ner fixed coil diameter, much greater than 24 cm. For 2a,

less than 16 cm, solutions continue but from figure 4c, one

can see that the number of turns on the inner fixed coils in- creases very rapidly without benefit; therefore it is impractical to consider a further decrease in 2a,. On the other hand, a larger 2a, decreases the total number of turns on

both coils. As an example, for 2a, = 20 cm, each of the outer fixed coils contains 3043 turns and the inner fixed

coils contain 20,169 turns. For 2a, = 24 cm, each of the outer fixed coils contains 2895 turns and the inner fixed

coils contain 13,518 turns. This corresponds to about 6000

m less wire. As for the current in these windings, 0.6 A pro132
duces a force of 0.049 N on the suspended coil when the suspended coil is energized with 10 Ampere-turns (At). Cor- respondingly 6 A generates a 0.49 N force and 60 A generates 4.9 N. Upon a current reversal of the 10 At in the suspended coil the counter-balancing mass required is 10, 102

, and 103 g respectively.

4. Coil geometry evaluation

Figures 5a, b, and c show the force gradient over the area

of interest where the maximum &.F0 on the suspended coil is approximately 10, 25, and 50 ppm respectively. The change

in the force is plotted based on the calculated force exerted on a "test" circular filament whose diameter is allowed to expand from 2a0 — 2 cm (76 cm) to 2a0 + 2 cm (80 cm). The

test filament is also displaced vertically about the mid-plane

(z = 0) to 3 cm above and below the mid-plane (z = +3 cm

and —3 cm). The radial distance, a0, on the plane corre- sponds to the radius of the suspended coiL The dotted line shows the path of the suspended coil as it travels vertically

about z = 0. Only the negative travel is shown since the

force gradients are symmetrical about the mid-plane. The

change in the peak LF0 occurs by changing only the spac- ings z, and z2 . The vertical force of course does decrease when the coil spacings are increased, about 0.6 percent

from AF0 = 10 ppm to AF0 = 50 ppm This decrease can be

then corrected by increasing the current in the fixed coils.

5. Current balance specifics

From figures 5a, b, and c, it can be seen that the extent of

available travel 2z0 of the suspended coil varies from about

4 cm for a AF0 of 10 ppm to about 6 cm for a AF0 of 50 ppm.

This distance must be accurately measured and pretty well

sets the scale of the apparatus to be built For a total vertical travel of the suspended coil from z = —3 cm to z = +3

cm, an uncertainty of 0.01 /im in the displacement will yield

a corresponding experimental uncertainty of 0.17 ppm for the case where AF0 is a maximum of 50 ppm

The dimensions of the apparatus used in all of the follow- ing evaluation calculations are: The diameter of the outer fixed coils (2a2 ) is 1.0 m. The diameter of the inner fixed coils (2a,) is 0.22 m. The diameter of the suspended coil (2ao) is 0.78 m. The separation of the outer fixed coils (z2) is 0.879 m. The separation of the inner fixed coils (z,) is 0.517 m. The current ratio or turns ratio {Nt of the inner fixed coil to that of the outer fixed coils is 5.54.

17.75 x 104 Ampere-turns on the outer fixed coils and 98.32 X 10* At on the inner fixed coils will produce a force on the

suspended coil that can be counter-balanced by a mass of 1 kg when the suspended coil contains 10 At. The magnetic field of the fixed coils will generate a one

volt d.c. signal across the suspended coil when the sus- pended coil containing 4081 turns traverses in the axial direction at a constant velocity of 0.5 mm/s. A current of

2.45 mA in such a suspended coil is required to produce a

force that will support a mass of 1 kg.

Obviously, the specifications just presented will require

that the fixed coils of the current balance be superconducting. A current of 60 A in the fixed coils will require that there be 2960 turns on the outer fixed coils and 16,390 turns on the inner fixed coils. The fixed coils are not very unusual

and should not be difficult to produce.

The suspended coil does present a problem if it is also to be at cryogenic temperatures. Since the balance will have to be at room temperature along with the standard mass, some

feed-through system would have to be devised which would

transfer the force on the suspended coil up to the balance

arm. The suspended coil cannot be in the liquid helium but

will have to be in a separate vessel and helium gas used to carry away the heat. An alternate method is to have the suspended coil in its own special dewar with that dewar

suspended from the balance beam. The first method allows

the laser interferometer optical components to be directly

mounted onto the coil forms. The index of refraction, however, will have to be continuously monitored. The second

method will have the interferometer optical components at- tached to the outside of the dewar. This will introduce an

additional uncertainty in the measurements of length. The

index of refraction however can be more easily monitored.

A third method is to have the suspended coil at room

temperature. Certainly a room temperature suspended coil greatly reduces the mechanical difficulties and should be

seriously considered.

The total length of wire needed on the suspended coil is 10,000 m With AWG 36 gauge oxygen free high conductivity copper wire, the resistance is 13.6 kQ. Measuring the generated d.c. emf with a 50 Hz band width limits the accu- racy to about 0.1 ppm due to Johnson noise. However inte- grating the voltage over a 120 s period limits the experimental accuracy to about 0.002 ppm When weighing, the cur- rent will produce about 90 mW of heating. This power will increase the temperature of the coil by less than 0.03 °C. Such a room temperature coil will have a mass of about 2.2 kg assuming the copper wire to be wound on a bobbin of low-expansion glass. Such a system is very appealing to start with and exhibits the possibility of reducing the uncertainty

in KA by an order of magnitude immediately. On the other hand, one can imagine the ultimate current balance having

the suspended coil at cryogenic temperatures. Such a sys- tem would compare the generated emf directly to the output

voltage of a Josephson junction at the 5 mV level Whereas

133
FIGURE 5. Figures a, 6, and c show the computedforce on a test current

filament, coaxial with the fixed coils, whose radius do is permitted to changefrom 38 to 40 cm.

This test filament is at the same time displaced vertically from the mid-plaDe(z = 0) to + and — 3

cm. At the mid-plane position the vertical force on the current filament is a maximum at ao = 39

cm. For off mid-plane positions, with Oq = 39 cm, the force increases to a maximum at i 0 (indicated by LF0 ). At this point the LF is at a minimum for changes in ao- The figure 5 mappings were made

for a 2d, of about 20 cm but the results are very similar for all 2a, which gave a useful solution.

134
ppm is given as the change in force from the midplane posi- tion for that particular misalignment. This misalignment

superimposes a linear force gradient on the ideal force gra- dient in the left column. The force at the mid-plane changes

a very small amount as indicated at the position z = 0 for the displacements of 0.004, etc. For the off mid-plane posi- tions of the suspended coil where z = 1,2 etc., the gradi- ents LF from the mid-plane force become quite asymmetric

rather rapidly. Fortunately, this misalignment is one which

can be measured and perhaps corrected for using tech- niques from the authors' previous experience in the gamma-p experiment (10, 11).

(2) Table lb shows the change in the vertical force on the suspended coil for the case where the inner fixed coil as- sembly is not coaxial with the outer and suspended coils (radial misalignment). The force gradients are symmetrical

about z = 0. The change in force at the mid-plane (z = 0) is with respect to the force for perfect alignment while the change in force for positions off the mid-plane are with

respect to the mid-plane force for that particular misalignment

TABLE la. The variation in the verticalforce on the suspended coilfor the case where either the inner or outerfixed coil assemblies are misaligned in the

axial direction. The quantity given is AF in ppm. The AF at z = 0 is referenced to the force for perfect alignment, while the &F at vertical displacements ofthe suspended coil is referenced to the force at the mid-planefor that particular misalignment

Displacement Misalignment (cm) in axial direction Z (cm) of suspended coil 0.000 0.004 0.008 0.012 0.016 0.020 + 4 ^12.7 -56.0 -69.4 -82.7 -96.1 -109.41 + 3 + 18.68 + 8.93 -0.817 -10.57 -20.32 -30.07 + 2 + 21.99 + 15.62 + 9.24 + 2.86 -3.51 -9.89 + 1 + 7.56 + 4.41 + 1.26 -1.89 -5.04 -8.19

0 0 -0.006 -0.025 -0.056 -0.10 -0.157 -1 + 7.56 + 10.716 + 13.87 + 17.02 + 20.17 + 23.32 -2 + 21.99 + 28.37 + 34.74 + 41.12 + 47.49 + 53.86 -3 + 18.68 + 28.42 + 38.17 + 47.91 + 57.65 + 67.39 -4 -42.7 -29.4 -16.0 -2.719 + 10.61 + 23.93

TABLE lb. The variation in the vertical-force on the suspended coil for the case where the innerfixed coil assembly is misaligned in the radial direction. The quantity given is AF in ppm. The AF at z = 0 is referenced to the force for perfect alignment, while the AF at vertical displacements ofthe suspended

coil is referenced to the force at the mid-planefor that particular misalignment.

Displacement Misalignment (cm) in radial direction Z (cm) of suspended coil 0.0 0.4 0.8 1.2 1.6 2.0 + 4 ^2.7 -36.9 -19.6 + 9.33 + 49.9 + 102.3 + 3 + 18.68 + 21.97 + 31.84 + 48.33 + 71.47 + 101.3 + 2 + 21.99 + 23.47 + 27.90 + 35.29 + 45.67 + 59.0 + 1 + 7.56 + 7.94 + 9.05 + 10.91 + 13.52 + 16.88 0 0 + 31.4 + 125 + 282 + 500 + 780

Note: AF is symmetrical about the midplane of the coil system.

the room temperature coil has an axial velocity of 0.5 mm/

to generate 1 V with a sampling time of 120 s, the all-cryo- genic system would have an axial velocity of 0.0025 mm/

and a sampling time of 24 x 103

s, assuming all other factors remained constant

6. Effect of Possible Coil Misalignments

It is generally easier to compute results for perfect coils ideally aligned than it is to produce such an assembly. The

following are the computed results for various misalign- ments.

(1) The change in the vertical force on the suspended coil for the case where the midplanes of the inner and outer coil pairs are not coincident turns out to be the most critical (see

table la). Table la shows the change in force in ppm on the suspended coil at the midplane z = 0 for misalignments in the axial direction of 0, 0.004, 0.008, 0.012, 0.016, and 0.020 cm. For the suspended coil at vertical positions of 1, 2, 3, and 4 cm away from the midplane, the change in force in

135
(3) Table lc shows the change in vertical force on the suspended coil for the case where the outer fixed coil as- sembly is not coaxial with the inner and suspended coils (radial misalignment). The change in force at the mid-plane

(z = 0) is with respect to the force for perfect alignment

while the change in force for positions off the mid-plane are with respect to the mid-plane force for that particular mis- alignment

(4) Table Id shows the change in vertical force on the suspended coil for the case where the suspended coil is not

coaxial with the inner and outer coils (radial misalignment).

It should be noted that the amount of misalignment given in tables lb- Id is 100 times greater than the axial misalignment of table la. The change in force on the suspended coil at the mid-plane for the outer fixed coils radially misaligned

is opposite the change in force for the case where the inner

fixed coils are radially misaligned. The change in force at the mid-plane (z = 0) is with respect to the force for perfect

alignment while the change in force for positions off the mid-plane are with respect to the mid-plane force for that particular misalignment. Even though the force change is substantial (some 30 ppm for a 4 mm displacement), there is very little change in the force gradient for the suspended

coil being off the mid-plane.

(5) Table 1 e shows the change in the vertical force on the suspended coil for the case where the suspended coil is tilted about the center of the system. The change in force at the mid-plane (z = 0) is with respect to the force for perfect

alignment while the change in force for positions off the mid-plane are with respect to the mid-plane force for that particular misalignment

(6) Table 1 f shows the change in the vertical force on the suspended coil for the case where the outer fixed coil as- sembly is tilted about the center of the system. The change

in force at the mid-plane (z = 0) is with respect to the force

for perfect alignment while the change in force for positions

off the mid-plane are with respect to the mid-plane force for that particular misalignment

(7) Table 1 g shows the change in the vertical force on the suspended coil for the case where the inner fixed coil as- sembly is tilted about the center of the system. The change

in force at the mid-plane (z — 0) is with respect to the force

for perfect alignment while the change in force for positions

off the mid-plane are with respect to the mid-plane force for that particular misalignment

Table lc. The variation in the vertical force on the suspended coil for the case where the outerfixed coil assembly is misaligned in the radial direction. The quantity given is AF in ppm. The AF at z = 0 is referenced to the forcefor perfect alignment, while the AF at vertical displacements ofthe suspended

coil is referenced to theforce at the mid-planefor that particular misalignment.

Displacement Misalignment (cm) in radial direction Z (cm) of suspended coil 0.0 0.4 0.8 1.2 1.6 2.0 + 4 -42.7 -42.3 -45.2 -48.4 -52.8 -58.5 + 3 + 18.68 + 18.32 + 17.26 + 15.48 + 12.99 + 9.78 + 2 + 21.99 + 21.83 + 21.36 + 20.57 + 19.47 + 18.04 + 1 + 7.56 + 7.53 + 7.41 + 7.21 + 6.93 + 6.58

0 0 -32.04 -128.2 -288.0 -513.0 -801.0

Note: AF is symmetrical about the midplane of the coil system.

Table Id. The variation in the verticalforce on the suspended coilfor the case where the suspended coil is misaligned in the radial direction. The quantity given is AF in ppm. The AF at z = 0 is referenced to the force for perfect alignment, while the AF at vertical displacements of the suspended coil is referenced to theforce at the mid-planefor that particular misalignment.

Displacement Misalignment (cm) in radial direction Z (cm) of suspended coil 0.0 0.4 0.8 1.2 1.6 2.0 + 4 -42.7 -37.6 -22.1 + 3.66 + 39.9 + 86.6 + 3 + 18.68 + 21.6 + 30.4 + 45.1 + 65.8 +92.5 + 2 + 21.99 + 23.3 + 27.3 + 33.9 + 43.2 + 55.1 + 1 + 7.56 + 7.90 + 8.89 + 10.5 + 12.9 + 15.9 0 0 -0.67 -2.75 -6.48 -12.3 -20.6

Note: AF is symmetrical about the midplane of the coil system.

136
(8) Table lh shows the change in the vertical force on the suspended coil for the case where both fixed coil assemblies

are tilted as a unit about the center of the system. The

change in force at the mid-plane (z = 0) is with respect to the force for perfect alignment while the change in force for positions off the mid-plane are with respect to the mid-plane

force for that particular misalignment

Of all the misalignments, the most critical is that where

the midplanes of the fixed coil pairs do not coincide. The

least critical alignment apparent in the vertical force on the suspended coil is where the suspended coil is either radially or angularly misaligned.

TABLE le. The variation in the vertical force on the suspended coil for the case where the suspended coil is tilted about the center of the system. The

quantity given is hF in ppm. The hF at z = 0 is referenced to the forcefor perfect alignment, while the AF at vertical displacements of the suspended coil

is referenced to the force at the mid-planefor that particular misalignment.

Displacement Tilt in degrees

Z(cm) of suspended coil 0.0 0.2 0.4 0.6 0.8 1.0 2.0 4.0 6.0 8.0

+ 4 -42.7 -43.1 -44.9 -47.9 -52.1 -57.5 -102.0 -283.0 + 3 + 18.68 + 18.28 + 17.28 + 15.56 + 13.20 + 10.12 -15.45 -118.0 -290.0 + 2 + 21.99 + 21.61 + 21.16 + 20.41 + 19.35 + 17.97 + 6.51 -39.4 -116.0 -225.0 + 1 + 7.56 + 7.41 + 7.30 + 7.11 + 6.85 + 6.50 + 3.62 -7.90 -27.2 -54.0

0 0 +0.0076 + 0.30 + 0.67 + 1.18 + 1.85 + 6.76 + 16.4 -3.47 -108.0

Note: AF is symmetrical about the midplane of the coil system.

Table 1 f. The variation in the verticalforce on the suspended coilfor the case where the outerfixed coil assembly is tilted about the center ofthe system. The quantity given is AF in ppm. The hF at z = 0 is referenced to the forcefor perfect alignment, while the AF at vertical displacements ofthe suspended

coil is referenced to theforce at the mid-planefor that particular misalignment.

Displacement Tilt in degrees Z (cm) of suspended coil 0.0 0.2 0.4 0.6 0.8 1.0

+ 4 -42.7 -42.6 -42.6 -42.7 -42.8 -42.9 + 3 + 18.68 + 18.56 + 18.54 + 18.49 + 18.51 + 18.43 + 2 + 21.99 + 21.71 + 21.78 + 21.69 + 21.76 + 21.72 + 1 + 7.56 + 7.40 + 7.45 + 7.38 + 7.45 + 7.44 0 0 -1.39 -5.75 -12.94 -23.08 -36.09

Note: AF is symmetrical about the midplane of the coil system.

Table 1 g. The variation in the vertical force on the suspended coil for the case where the inner fixed coil is tilted about the center of the system. The

quantity given is AF in ppm. The AF at z = 0 is referenced to the forcefor perfect alignment, while the AF at vertical displacements ofthe suspended coil

is referenced to the force at the mid-planefor that particular misalignment.

Displacement Tilt in degrees

Z(cm) of suspended coil 0.0 0.2 0.4 0.6 0.8 1.0

+ 4 -42.7 -43.1 -44.9 -48.0 -52.0 -57.4 + 3 + 18.68 + 18.28 + 16.80 + 15.87 + 18.08 + 10.53 + 2 + 21.99 + 21.93 + 21.28 + 20.54 + 19.65 + 18.42 + 1 + 7.56 + 7.34 + 7.24 + 7.09 + 6.88 + 6.82 0 0 -4.50 -18.1 -41.1 -73.2 -114.7

Note: AF is symmetrical about the midplane of the coil system.

137
Table lh. The variation in the vertical force on the suspended coil for the case where the entire fixed coil assembly is tilted about the center of the system. The quantity given is A/7 in ppm. The hF at z = 0 is referenced to the force for perfect alignment, while the LF at vertical displacements of the suspended coil is referenced to the force at the mid-planefor that particular misalignment.

Displacement Tilt in degrees Z (cm) of suspended coil 0.0 0.2 0.4 0.6 0.8 1.0

+ 4 -42.7 -43.2 -44.9 -48.1 -52.3 -57.8 + 3 + 18.68 + 18.20 + 16.71 + 15.73 + 12.% + 10.33 + 2 + 21.99 + 21.87 + 21.29 + 20.46 + 19.64 + 18.38 + 1 + 7.56 + 7.30 + 7.25 + 7.03 + 6.88 + 6.83

0 0 -5.90 -23.86 -54.08 -96.3 -150.7

Note: AF is symmetrical about the midplane of the coil system.

7. Non Vertical Forces

Perhaps even more important than the change in vertical forces for coil misalignment, is the unmeasured horizontal

forces acting on the suspended coil. In the calibrating part

(the measuring of the generated emf) of the experiment, a horizontal force translates into an error emf if there is any

horizontal velocity. This "horizontal" emf adds directly to the vertical emf and becomes an error in the force calibra- tion unless it is properly accounted for. Table 2a shows the horizontal force on the suspended coil for a misalignment in the radial direction of the suspended

coil and for vertical positions of the suspended coil up to z — 4 cm from the mid-plane of the system. The horizontal

force is an odd function about the mid-plane. The forces given here are in parts per million (ppm) of the vertical

force. The sign indicates direction Le. if negative, the force

is away from the central position, positive is a restoring

force. Of course if the direction of current is reversed in the suspended coil then the direction of all forces is reversed.

Table 2b is a tabulation of horizontal forces for the case where the suspended coil is tilted. Results are shown for tilts up to 2 degrees. The force is perpendicular to the axis of rotation of the coil. The horizontal force is given in ppm of

the vertical force for perfect alignment

The classical current balance configuration has the characteristic that when there is just angular (tilt) misalignment,

there is no torque on the suspended coil at the mid-plane.

Table 3a summarizes the torques present on the suspended

coil when the only misalignment is a tilt over the coil range

of vertical displacement of the suspended coil from z = +4 cm to —4 cm. The values given are in units of length (/im). The torque is normalized to the vertical force of the perfectly aligned system.

Table 3b shows the torque for the case where the inner

fixed coil pair is tilted. Table 3c shows the torque for the case where the outer fixed coil pair is tilted. These torques

are opposite and tend to cancel when it is the entire fixed

coil assembly that is tilted.

Table 2a. Ratio of horizontal force to vertical force (in ppm) on the suspended coil for the case where the suspended coil is misaligned in the radial

direction.

Displacement Misalignment (cm) in the radial direction Z(cm) of suspended coil 0.0 0.1 0.2 0.3 0.4 0.5 0.8 1.2 1.6 2.0

+ 4 0.0 -10.7 -21.2 -31.4 -41.1 -49.9 -70.2 -75.2 -43.7 + 37.5 + 3 0.0 -2.4 -4.7 -6.7 -8.2 -9.2 -7.2 -12.5 + 60.4 + 146.5 + 2 0.0 + 1.1 + 2.3 +3.7 + 5.4 + 7.5 + 17.2 + 41.6 + 85.2 + 154.8 + 1 0.0 + 1.4 + 2.8 + 4.4 + 6.0 + 7.9 + 15.2 + 30.9 + 56.2 + 94.7 0 0.0 0.0 0.0 0.0 0.0 0.0 0.0 0.0 0.0 0.0 -1 0.0 -1.4 -2.8 -4.4 -6.0 -7.9 -15.2 -30.9 -56.2 -94.7 -2 0.0 -1.1 -2.3 -3.7 -5.4 -7.5 -17.2 -41.6 -85.2 -154.8 -3 0.0 + 2.4 + 4.7 + 6.7 + 8.2 + 9.2 + 7.2 + 12.5 -60.4 -146.5 -4 0.0 + 10.7 + 21.2 + 31.4 + 41.1 + 49.9 + 70.2 + 75.2 + 43.7 -37.5

138
TABLE 2b. Ratio of horizontal force to vertical force (in ppm) on the suspended coilfor the case where the suspended coil is tilted about the center of the

system.

Displacement Tilt in degrees Z (cm) of suspended coil 0.0 0.4 0.8 1.2 1.6 2.0

+ 4 0.0 + 3.9 + 8.1 + 12.4 + 17.2 + 22.7 + 3 0.0 + 0.90 + 1.9 + 3.0 + 4.4 + 6.1 + 2 0.0 -0.03 -0.02 + 0.06 + 0.25 + 0.58 + 1 0.0 -0.09 -0.17 -0.23 -0.27 -0.27

0 0.0 -0.03 -0.05 -0.08 -0.11 -0.14

Note: The force ratio is symmetrical about the midplane of the coil system.

TABLE 3a. Ratio of torque to vertical force acting on suspended coil (unit of\xm) for the case where the suspended coil is tilted about the center of the

system.

Displacement Tilt in degrees Z (cm) of suspended coil 0.0 0.2 0.4 0.6 0.8 2.0 4.0

+ 4 0.0 -13.8 -27.6 -41.4 -55.4 -135.8 -264.1 + 3 0.0 -10.5 -20.9 -31.4 -41.9 -103.0 -200.5 + 2 0.0 -7.04 -14.1 -21.1 -28.1 -69.2 -134.8 + 1 0.0 -3.54 -7.07 -10.6 -14.1 -34.8 -67.8 0 0.0 0.0 0.0 0.0 0.0 + 0.0 0.0 -1 0.0 + 3.54 + 7.07 + 10.6 + 14.1 + 34.8 + 67.8 -2 0.0 + 7.04 + 14.1 + 21.1 + 28.1 + 69.2 + 134.8 -3 0.0 + 10.5 + 20.9 + 31.4 + 41.9 + 103.0 + 200.5 -4 0.0 + 13.8 + 27.6 + 41.4 + 55.4 + 135.8 + 264.1

TABLE 3b. Ratio oftorque to verticalforce acting on suspended coil (unit of\im)for the case where the innerfixed coil is tilted

Displacement Z (cm) of suspended coil

Tilt in degrees

0.0 2.0 4.0 6.0 8.0

+ 4 0.0 + 34.0 + 69.0 + 106.4 + 147.2 + 3 0.0 + 24.1 + 49.0 + 75.6 + 104.8 + 2 0.0 + 15.4 + 31.3 + 48.4 + 67.2 + 1 0.0 + 7.5 + 15.2 + 23.6 + 32.8 0 0.0 0.0 0.0 0.0 0.0 -1 0.0 -7.5 -15.2 - 23.6 -32.8 -2 0.0 -15.4 -31.3 - 48.4 -67.2 -3 0.0 -24.1 -49.0 -75.6 -104.8 -4 0.0 -34.0 -69.0 -106.4 -147.2

Table 3c. Ratio oftorque to verticalforce acting on suspended coil (in units offim)for the case where the outerfixed coil is tilted

Displacement Tilt in degrees Z (cm) of suspended coil 0.0 2.0 4.0 6.0 ao

+ 4 0.0 -169.8 -333.2 -483.6 -613.6 + 3 0.0 -127.1 -249.5 -362.2 -460.1 + 2 0.0 -84.6 -166.1 -241.3 -306.6 + 1 0.0 -42.3 -83.0 -120.6 -153.3 0 0.0 0.0 0.0 0.0 0.0 -1 0.0 + 42.3 + 83.0 + 120.6 + 153.3 -2 0.0 + 84.6 + 166.1 + 241.3 + 306.6 -3 0.0 + 127.1 + 249.5 + 362.2 + 460.1 -4 0.0 + 169.8 + 333.2 + 483.6 + 613.6

139
Table 3d shows the torque on the suspended coil when

there is a misalignment in the radial direction between the

fixed coil assembly and the suspended coiL This torque is by far the largest of all the other torques calculated. The

torque is relatively constant over the expected vertical posi- tions of the suspended coil and increases fairly linearly with

increased misalignment

Most of the effects of coil misalignment are small, but

they will have to be monitored and corrections will have to be applied. For the case of vertical forces, if perfect alignment is not possible, the experimental accuracy will be

unaffected, as long as the misalignment does not change

during the course of operation. The chance for a change in alignment can develop during the measurement of voltage.

TABLE 3d. Ratio of torque to vertical force acting on suspended coil (in units of\im) for the case where the suspended coil is misaligned in the radial

direction.

Displacement Tilt in degrees Z (cm) of suspended coil 0.0 0.2 0.4 0.6 0.8 1.0

+ 4 0. 849 1696 2546 3396 4248 + 3 0 849 1698 2548 3398 4250 + 2 0 849 1698 2548 3399 4251 + 1 0 849 1698 2548 3399 4250

0 0 849 1698 2548 3398 4250

Note: AF is symmetrical about the midplane of the coil system.

In this instance, the suspended coil will be changing its axial position at a uniform velocity with respect to the fixed

coils. This action can occur by displacing the fixed coils, by

displacing the suspended coil through movement of the entire balance, or by allowing the balance to swing. Either of

the three methods will require a positive servo system to either keep the velocity constant or the generated emf con- stant Certainly, the swinging of the balance is the most at- tractive since it requires the least effort The swinging of

the balance however naturally produces a misalignment in the radial direction of the suspended coil when it travels from +z0 to —

z

0. Depending upon the length of the balance

beam, though, this displacement can be kept smalL For a balance beam with a half length of 40 cm, the maximum

radial displacement is about 0.1 1 cm. From Table 2a, it can

be found that the horizontal force is about 2.6 ppm of the

vertical force. This horizontal force results in an additional displacement of less than 1 Lim when the vertical distance

from the knife edge to the suspended coil is 1 m. The hori- zontal force produced is a negligible error in the vertical force measurement However, this horizontal force translates into a generated emf into the suspended coil during

the calibration part of the experiment and yet does not add

to the vertical force. This is a potential systematic error if not properly taken into account The maximum calculated horizontal velocity is 3.75 x

10"3 cm/s as compared to the vertical velocity of 5 x 10

"2 cm/s. This will increase or decrease the emf measured at the end points by 0.195 /iV in one volt If the initial velocity

is negative at z — +3 and the Aemf is positive then the Aemf

will also be positive at z = —3. Therefore a systematic error

will result and a correction must be applied. This correction can be calculated from a measurement of the horizontal

force.

8. Conclusion

The first part of the experiment involves calibrating the

force balance through the measurement of a d.c. voltage

and a velocity. Both of these measurements are difficult but

possible at the 0.1 ppm level of uncertainty.

The second part of the experiment "weighing the force", will be relatively easy. Considering that the mass of

the suspended coil will be approximately 2.2 kg and the vertical electromagnetic force will be counter-balanced by mass

of 1.0 kg, a weighing to 5 x 10"' (16 /ig) will result in an uncertainty of 0.02 ppm in the force measurement While a

guest worker at NBS, Mr. H. Nakamura of the Electrotechnical Laboratory, Japan developed a very good servo-system

for the NBS current balance [10]. Mr. Nakamura's contribution will certainly allow the force to be weighed to 0.02 ppm

or better. We hope that the apparatus outlined above (resembling

fig. 6) will resolve the confusion that now exists in present

determinations of KA . The undertaking is a large scale ex- periment with much apparatus to be constructed. The exact time required for achieving the ultimate accuracy in the ex- periment is not easily determinable, nor is the ultimate ac140
General Appearance of Compensated Current Balance

(not to scale)

FIGURE 6. The general appearance of the compensated current balance

is shown.

The mus on the left scale pan of the balance is sufficient to counterbalance the mass of the sus- pended coil plus the downward vertical force exerted on the suspended coil When the current in the suspended coil is reversed, then a mass M is added to the right scale pan to offset the upward vertical force. Thus the mass M is sufficient to counterbalance the upward vertical force plus the negative of the downward vertical force.

curacy itself. However, with reasonable support, it may be

expected that a value with an uncertainty in the neighborhood of a few tenths ppm is possible within a few years. The new ampere measurement will not only help resolve the discrepancies which now exist among several fundamental constant determinations, but will also enable the present

value of 2e/h used for volt monitoring applications to be

brought closer to its absolute value in terms of mass, length,

and time. In the future, the sensitivity of the apparatus can

be used to monitor the stability of the kilogram with respect

to time, by utilizing the stability of the Josephson voltage

standard and the calculable capacitor. Since the force developed is sufficient to counterbalance one kilogram with

only 2 mA in the suspended coiL the apparatus can be used

to calibrate other mass standards in terms of the kilogram

by merely determining a current ratio. Even though the absolute ampere experiment as outlined above is an ambitious project, one must consider that the authors propose to measure the ampere to an accuracy 50

times better than it has ever been measured, and that absolute ampere determinations have a 100 year history of ef- fort Advances do not come easily.

The authors wish to thank Mr. Robert E. Kleimann and

Mrs. Sheila A. Taylor for their assistance in writing the computer programs and in the operation of the computer.

9. Appendix A

The derivations leading to equation 8 can also be

described in an integral form. This integral approach is very useful because the quantities that come into the final result are more easily measured in the laboratory.

The force in the z direction between two coils of a mutual

inductor is given by

Fz = hUdMl2/dz) (Al)

The integral of this force from an initial position z, to a

final position zf

is: £FJz = hh[Ml2{zf) - MM). (A2)

The emf generated in coil 1 when a current 72 is in coil 2 is

Ut) = -h(dM12 /dt) (A3)

which can be integrated from time to t2:

J

2 e{t)dt = -hWM-MMl (A4)

Now if i, is the time the moving coil passes the position z, and < 2 the time the coil pass zf then we can divide eq (A2) by

eq(A4)

A = (f

Zf Fz dz/f

2 ei(t)dt) (A5)

This is the integral form of eq (3) and can be written explicitly in terms of NBS units so that eq(8) becomes

KA = (ANBS/A) =

J m(z)gdz

INBs(QNBs/Q)f'eNBs(t)dt

(A6)

The f'm(z)g dz is computed from many static weighings

z • made along the path from z,- to z, with a current INBS in the balance coil. The/

2 £Ni>s(t)dt is measured in a separate ex- periment when the* coil is moved from position z, to z, in a time ti — t2 at a nearly constant velocity and the current INBS

is not flowing in coil 1. The geometry and the current of coil 2 must remain unchanged during all these measurement or an appropriate correction must be made.

141
10. Appendix B

An interesting variation on this method for measuring the ampere occurs if one restricts the moveable coil to a single, rotational degree of freedom. Such a rotatable coil placed in the nearly uniform field produced by an external circuit forms the basis of the classic electrodynamometer or Pellat balance [3] The torque on the rotatable coil is given by

1\6) = IJ2(dM12 /d6) (Bl)

where /, and I2 are the currents in the external and rotat- able circuits respectively, M12 is the mutual inductance and

6 is the rotation angle. The integral of this torque in moving

the coil betwen two angles is then

W=ff T\d)dd = hh f'dMl2 = IJ2 [M12 (6f

) - M12(6,)] (B2)

If the coil is now allowed to rotate with its circuit open,

the magnitude of the emf generated is

= UdMxl/dt) (B3)

Integrating this emf over time one obtains

j\(t)dt = I, f' dM12 = h[MM ~ MM] (B4)

If <, and t 2 are the times when the coil is at angles 0, and Qf of eq (B2) we may divide eq (B2) we may divide eq (B2) by eq

(B4) to obtain

/, = J

f 7ld)d6/j\(t)dt (B5)

6

i 'l Since the numerator of eq (B5) is just the work done by

the electrical torque in rotating the coil between 0, and 0,, it can be replaced with any equivalent expression for this work. In particular, this work is equal to the work done by a

vertical external force applied at the end of a balance beam

attached to the moveable coil when that external force just

balances the electrical force. This force is what can be

measured experimentally by weighing. Expressed this way

eq(B5) becomes

/, = fFz(z)dz/J

2£(t)dt, (B6)

'i where z, and z, are the vertical positions of the end of the balance arm at angles 0, and 0,. This is identical to eq (A5)

and leads directly to eq(A6) forKA . A major disadvantage of the method described in this ap- pendix is that the force and the emf depend directly on the area of the moveable coil, in contrast to the geometry de- scribed in the body of the text, where they are essentially

independent of the area of the moveable coil. Because of

this, temperature control problems are more critical for the rotating coil. On the other hand, the rotary coil has the ad- vantage of being constrained to move with one degree of freedom. As a result problems of non-vertical forces and

torques which must be considered in the coil system de- scribed in the text do not affect the rotating coil. Since no

displacements other than rotation are allowed, no emfs are generated and no work is done other than by the forces explicity considered.

11. Appendix C

The true current balance field coils will have windings of

finite cross section. Below, results and specifications have

been tabulated for field coils which must provide enough

radial magnetic field to exert a vertical force of 4.9 N on a

circular filament of 78 cm diameter containing a 10 ampere

current The outer field coil consists of a bundle of square cross section containing 54 x 54 of 0.0508 cm diameter conductors. The mean diameter of the coil is 100 cm. The inner field coil consists of a bundle of rectangular

cross section containing 115 (radial) x 116 (axial) of 0.0508 cm diameter conductors. The mean diameter of the coil is 24 cm.

Table CI.

z2 (cm) z, (cm) Uh AF0 (ppm) 2Az0 (cm)

86.1186 50.6786 59.1834593 1.0021216 7.1 3.5 86.2328 50.7328 59.2713175 1.0019726 12.3 3.9 86.3470 50.7848 59.3612670 1.0017318 19.2 4.5 86.5090 50.8712 59.4762047 1.0019700 33.2 5.1 86.5710 50.9024 59.5219867 1.0019812 39.2 5.4 86.6292 50.9314 59.5652223 1.0019806 45.4 5.6 86.7336 50.9832 59.6429316 1.0019745 58.0 6.2 86.8282 51.0302 59.7132118 1.0019770 70 6.3

142
12. Appendix D Table D-l

The coincidence of the mid-planes of each of the two coil pairs is the most critical alignment. A misalignment, in the

axial direction, of the mid planes by as much as 0.004 cm

begins to introduce an undesirable linear gradient in the

force along the z-direction. This misalignment could be cor- rectable by mechanically adjusting either coil pair along the

axis. However figure D-l shows the positions of correction

coils which provides the ability to effectively "reposition"

the large field coils by + or — 0.2 cm in the axial or z-direc- tion electrically. Table D-l gives the ampere- turns, as a per- cent of the ampere-turns of the main coil required to effectively reposition one of the large diameter field coils in the

axial direction. Each correction coil has the same number of

turns. Electrically they are connected in opposition.

MAIN COIL #

CORRECTION COILS ® Q

Figure D-l.

Coil misalignment in cm

correction coil ampere-turns in percent of main ampere-turns -0.20 -3.04519

-0.15 -2.28639

-0.10 -1.52593

-0.05 -0.76380

+O.05 +0.76547

+0.10 +1.53262

+0.15 +2.30145

+0.20 +3.07197

13. References

[I] Driscoll, R. L., and Cutkosky, R. D. Measurement of Current with the National Bureau of Standards Current Balance, J. Res. Nat. Bur. Stand. (U.S.), Vol. 60, No. 4, 297-305 (April 1958).

[2] Curtis, H. L., Curtis, R. W., and Critchfield, C. L. An Absolute Deter- mination of the Ampere, Using Improved Coils, J. Res. Nat. Bur. Stand. (U.S.), Vol. 22, 485-517 (May 1939).

[3] Driscoll, R. L. Measurement of Current with a Pellat-Type Electro- dynamometer, J. Res. Nat. Bur. Stand. (U.S.), Vol. 60, No. 4, 287-

296 (April 1958).

[4] Driscoll, R. L, and Olsen, P. T. report to Co mite Consul tatif d'Electricite, Comite International des Poids at Measures, 12th Session,

(1968).

[5] Taylor, B. N. Is the Present Realization of the Absolute Ampere in Error, Metrologial2,81 (1976).

[6] Kibble, 8. P., and Robinson, I. Feasibility Study for a Moving Coil Apparatus to Relate the Electrical and Mechanical SI Units, NPL

Report DES No. 40,(1977).

[7] Kibble, B. P., A Measurement of the Gyromagnetic Ratio of the Proton

by the Strong Field Method, Atomic Masses and Fundamental

Constants 5, ed. by J. H. Sanders and A. H. Wapstra (Plenum Publishing Corp., New York, 1976) p. 545.

[8] Cutkosky, R. D., New NBS Measurements of the Absolute Farad and Ohm, IEEE Trans. Instrum. Meas. IM-23, 305, (1974).

[9] Snow, C, Formulas for Computing Capacitance and Inductance, NBS

Circular 544 (1954).

[10] Williams, E. R., and Olsen, P. T., A Non-contacting Magentic Pick-up Probe for Measuring the Pitch of a Precision Solenoid, IEEE

Trans. Instrum. Meas., IM-21, 376(1972).

[II] Olsen, P. T., and Williams, E. R., A More Accurate Determination of

yp Through Improved Dimensional Measurement Techniques, IEEE Trans. Instrum. Meas., IM-23, 302 (1974).

[12] Nakamura, H., A Serv-Controlled Balance for the Absolute Ampere

Determination, Jap. J. Appl. Phys. 17, 1397, (1978).

143
Reprinted by permission from IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT

Vol. IM-23, No. 4, December 1974, pp. 305-309 Copyright 1975 by the Institute of Electrical and Electronics Engineers, Inc. PRINTED IN THE U.S.A.

New NBS Measurements of the Absolute Farad and Ohm

ROBERT D. CUTKOSKY, fellow, ieee

Abstract—A recently completed calculable cross capacitor in conjunction with a previously described collection of ac and dc bridges has made possible a highly accurate measurement of the farad and

the ohm. The cross capacitor and its auxiliary equipment, as well

as those components of the measurement system which have not been covered in prior publications, are described in detail. The

measurements indicate that the National Bureau of Standards

(NBS) unit of capacitance is given by FNBs = 1 F + 1.787 ^F, and

that the NBS unit of resistance is given by 0NB8 = 1 £2 — 0.819 nQ. INTRODUCTION

FOLLOWING the theoretical and experimental development of the calculable cross capacitor by

Thompson and Lampard of the National Standards

Laboratory of Australia [1], a research effort was initiated

Manuscript received July 3, 1974; revised August 20, 1974. This paper is a contribution of the National Bureau of Standards and is not subject to copyright.

The author is with the Electricity Division, National Bureau of Standards, Washington, D. C. 20234.

at the National Bureau of Standards (NBS) to utilize these principles in an absolute measurement of the farad

and the ohm. The initial NBS effort in this field [2] was

completed in 1960 and provided an absolute measurement

of the ohm with an estimated uncertainty of about 2 parts

in 106. These measurements provided the basis for the

present NBS unit of capacitance, F\ Bs- The NBS unit of resistance Qnbs ^'as not adjusted at that time, because the

unit in existence since 194S was found to be substantially

correct. The activities of our laboratory since 1960 have

been directed toward improving the equipment and techniques which were used in the initial effort with the

objective of completing an absolute ohm measurement

with an uncertainty on the order of a few parts in 108

. ORGANIZATION OF MEASUREMENT SYSTEM

The starting point in the NBS absolute ohm measure- ment is a O.o-pF calculable cross capacitor, the capacitance

144
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1974

of which can be determined from a measurement of its length. This instrument is used to measure the capacitance

of a transportable 10-pF reference capacitor which can be

carried from the basement room containing the cross ca- pacitor to a large second floor . laboratory containing the

other components of the measurement system.

The second floor laboratory contains the bank of 10-pF

reference capacitors that maintain the NBS unit of capacitance. An ac bridge with a nominal voltage ratio of 10:1

is used in two stages to measure two 1000-pF capacitors.

These capacitors are used in turn as two arms of a special

frequency-dependent bridge called a quadrature bridge

for measuring the ac resistances of two 105-fi resistors. These resistors are then compared with a transportable

resistor through the use of a 100:1 ac bridge that

utilizes the 10:1 voltage ratio bridge just cited and a 10:1 current comparator [3]. The transportable 1000-ft resistor is then carried to the

laboratory in which the NBS bank of 1-Q reference resistors

is maintained. The stepup from the level to the 1000-Q

transportable resistor is performed in this laboratory using

dc techniques with the aid of Hamon dividers and a dc

comparator.

Essential special auxiliary equipment includes a low

noise parametric preamplifier, instruments for measuring

the voltage dependence of capacitors and other components, a transformer calibrator, and all of the equipment

necessary for determing the frequency dependence of the

cross capacitor and of the transportable 1000-0 resistor. Except for the cross capacitor itself and equipment related to its use, all of the special instruments and techniques

used in this work have been described in a series of publications dating back to 1962 [3]-[7]. The attention of

this paper is accordingly concentrated on the construction and operation of the cross capacitor and on those aspects

of the measurement system which have not already been

adequately covered.CROSS CAPACITOR

The NBS cross capacitor utilizes the now classic geometry which was first exploited by Clothier [8]. Our system

contains four vertical cylindrical electrodes with diameters

of 6.35 cm which were lapped to an accuracy of ± 0.1 nm

and which are located in a square array with clearances of 3.60 mm between adjacent electrodes. A grounded cylindrical shield with an inside diameter of 18.16 cm surrounds

the four electrodes. A movable shield rod with a diameter

of 2.72 cm is partially inserted between the electrodes from

the top and is located horizontally by polytetraflouroethylene rings which are placed around the shield rod and

which lightly touch all four electrodes. This shield rod may

by translated vertically by means of a pulley and counter- weight system to vary the cross capacitance of the instru- ment in a controlled manner. The lower end of the shield rod is composed of an Invar fitting which is terminated

with a vertical tube 22.4 mm long with an outside diameter

of 10.0 mm. The geometry of this tube was chosen to eliminate the first-order dependence of capacitance upon

electrode taper and angular misalignment which could

otherwise seriously affect the measurements. An optical flat is mounted inside the Invar fitting just

above the misalignment compensation tube. A similar Invar fitting with a second optical flat is mounted just

above the lower ends of the electrodes. The lower fitting can be adjusted either mechanically or electrically by means of a piezoelectric transducer (PZT) to make the

flats precisely parallel. The PZT can also be used to dis- place the lower flat vertically over a range slightly larger

than 1 fringe. Provision has been made to dither the lower

flat over a range of a few hundredths of a fringe at a fre- quency of 10 Hz. The two flats form the plates of a FabryPerot interferometer which is iiluminated by a Lamb-dip

stabilized helium-neon laser. The optical system contains a polarizer and quarter-wave plate to minimize the amount

of light returning to the laser, a beam expanding telescope

to fill the 7-mm-diameter Fabry-Perot resonator, a 50-cm

focal length lens to focus the interference fringes, and an

adjustable pinhole at the focal plane of the lens to select the light from the center of the fringe pattern and to allow

it to strike a phototransistor. The 10-Hz component of the

phototransistor output resulting from dithering the lower

flat is amplified and detected with a phase sensitive detector. The detector output is amplified and fed back to the PZT to lock the interferometer on the center of a fringe.

The shield rod contains two cylindrical sections 7 mm

long which are electrically insulated from the rest of the

shield rod. These sections are of the same diameter as the

shield rod and are connected via sliding contacts to sep- arate terminals on the housing of the capacitor. One of these sections is located a few millimeters above the Invar

fitting containing the upper flat, and the other is about

10 cm farther up. These sections make it possible to monitor variations in the interelectrode spacing and in the angle

between the shield rod and the electrode axis as the shield rod is displaced through its range of travel through measurements of the direct capacitance between each cylindrical section and each capacitor electrode. These sections

are shorted to the case of the capacitor when cross-capaci- tance measurements are being made.

The cross capacitor becomes slightly nonlinear when

the shield rod is set to produce a cross capacitance smaller than 0.2 pF, and has a maximum capacitance of 0.7 pF.

Within this normal operating range of 0.5 pF, only the expected nonlinearity at the 0.2-pF end of the range has

been found. The capacitance nonlinearity at the 0.2-pF

end of its range was determined by comparing the rates of change of capacitance with shield rod displacement, with

the shield rod set to yield various cross capacitances down

to 0.06 pF. The function relating departure from linearity to normal capacitance was found to be exponential and

indicated that at the 0.2-pF setting the nonlinearity

amounted to 7.0 X 10-9 pF. A correction of 14 parts in

in 109 was accordly applied to the 0.5-pF capacitance

difference. The uncertainty in cross capacitance caused by imprecise mechanical construction and alignment is estimated

145
cutkosky: NUS MEASUREMENTS

to be smaller than 1 part in 108

. A correction of 3 parts

in 109 to account for a slight error in the laser beam di- rection was applied. The wavelength of our present laser

is unfortunately not completely stable, and at present

limits our capacitance accuracy. The capacitor is normally

evacuated to a sufficiently small pressure so that correction

for changes either in the laser wavelenght or in the di- electric constant of the capacitor is unnecessary. The distributed capacitances and inductances in the

leads and electrodes of the calculable capacitor result in a small frequency dependence. The distributions of these

parasitic impedances were determined by direct measure- ments coupled with some estimates of end effects. The

total correction for these internal loading effects including the capacitive loading correction for the transformer used

to compare the calculable capacitor with the 10-pF trans- portable reference capacitor was found to be +9 parts

in 109 at 1.592 Hz. ALTERNATING CURRENT MEASUREMENTS

The capacitance of the calculable capacitor is compared

with that of a transportable 10-pF reference capacitor in a bridge containing a special transformer which supplies

200 V at 1592 Hz to the capacitor. When the calculable capacitor is set to yield 0.2 pF, the reference capacitor is connected to a tap supplying 4 V, and when it is set to

yield 0.7 pF the reference capacitor is connected to a tap

supplying 14 V. The critical ratios of this transformer were measured using larger fixed capacitors which can be ac- curately calibrated through the use of a previously described 10:1 voltage ratio bridge [3]. The open circuit ratio correction for the special transformer was found to be — 71 parts in 109

. The remaining features of the bridge

used for comparing the cross capacitor with the 10-pF

reference capacitor and in determining the fringe number

are similar to those described by Clothier [8]. The impedance of the transportable 10-pF reference capacitor at 1592 Hz is compared with the impedance of a

1000-ft transportable resistor using a set of bridges de- scribed in an earlier paper [3]. The four-pair 10:1 voltage

bridge was calibrated through the use of a set of eleven 10-pF capacitors by means of the permutation method [9],

but modifications were made so that better advantage

could be taken of the four-pair configuration. The voltage

dependencies of all of the capacitors involved in the stepup and in the calibrations were measured on an absolute

basis using a new version of the system described by

Shields [10]. The only capacitors for which the voltage

dependence corrections were larger than 1 part in 109 were two 1000-pF capacitors used to calibrate a 10:1 ac current comparator. This comparator was part of a 100:1 bridge

used for the ac measurement of the 1000-ft transportable

resistor. Two different pairs of capacitors were used for this calibration. The voltage dependencies of the capacitors over the 20-V to 200-V range varied from 82 parts in 109 to 154 parts in 109

, but after the corrections were

applied, all of the measurements of the current comparator

ratio agreed within 2 parts in 109

.

TRANSPORTABLE RESISTOR

The 1000-O transportable resistor consists of nine 1000-ft Evanohm resistors wound on mica cards. The individual cards are connected in series parallel and mounted in a

sealed metal can nearly filled with oil. The can is in turn, immersed in a temperature-regulated oil bath. This re- sistor is measured with both ac current and with dc current, and it is necessary to correct for any ac-dc resistance

difference which may exist. Two special coaxial resistors, one of 100 ft and one of 1000 ft, were constructed to investi- gate this possibility [11]. Each of these resistors consists of a single length of Evanohm wire about 30 cm long and of suitable diameter mounted along the axis of an oil-filled tubular return conductor with a diameter of 5 cm. Shielded

current and potential terminals were provided at each end. Complete calculations of the field distributions and eddy

currents were made to allow an assessment of the frequency

dependencies of the two resistors. A comparison of the

ratios of the two resistors at 1592 Hz and at 15920 Hz

showed close consistency with the calculations, both in resistance and in phase angle. On the basis of the calculations, the 1000-ft coaxial resistor was assigned a resistance

difference from dc to 1592 Hz of S parts in 1010 which is not significant, but which does not include a possible dc

error attributable to the Peltier effect [12], which could

arise from the use of spot welding to fasten the resistance wire to its terminals. A microscopic examination of the welded junction showed a discolored section of Evanohm

wire extending out from the weld for a distance of about

0.2 mm. It was subsequently determined that annealing

a piece of Evanohm wire substantially changes its thermoelectric coefficient, so that a thermoelectric junction could

exist 0.2 mm from each end of the resistance wire, possibly

causing the dc resistance to be abnormally high due to the

Peltier effect. A second special 1000-ft resistor was built to determine

whether or not the Peltier effect in the coaxial 1000-ft

resistor was significant. This resistor consists of a length

of resistance wire about 3 m long folded twice to form a

four-wire cage 75 cm long. The wires are separated by 1 cm

throughout the length of the system and produce a quadrupole field, which is not strongly coupled to the metal

supports and shield structure. Current and potential terminals for each end of the resistor consist of pieces of 0.07-mm diameter copper wire 1 cm long soldered to heavier copper lead wires with low thermal solder. The

0.07-mm wires were placed parallel to each other and about

3 mm apart on a beryllium oxide heat sink. The resistance wire was placed over the copper wires and perpendicular

to them and clamped in place with a second beryllium

oxide heat sink. The contact resistance of this purely

mechanical connection was about 0.1 ft. The resistor was

easily trimmed to ± 100 ppm by releasing the clamping

pressure and moving the resistance wire as required. The

finished resistor was immersed in an oil bath for temperature control. The quadrupole resistor was not expected to exhibit an

146
IEEE TRANSACTION'S ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1974

appreciable Peltier effect because of the absence of thermoelectric junctions in the portions of the resistor common to the current and potential circuits. A verification of the absence of Peltier errors was made by connecting a nanovoltmeter between a normally open extension of the Evanohm resistance wire and the adjacent clamped junction. A current was then passed through the 1000-0 section

of the resistor via the innermost of the four clamped junctions. Reversing the sign of the current had no effect on

the nanovoltmeter reading. Some eddy current errors could exist, but this would cause an apparent increase of resistance with frequency.

The coaxial 1000-0 resistor is believed to be substantially

free of eddy current errors, but if it were troubled with the

Peltier effect, its resistance would appear to decrease with

frequency. Measurements of the ratio of the two special 1000 0 resistors at dc and at 1592 Hz agreed within 5 parts

in 109

, which is interpreted as indicating that neither

resistor has a significant frequency dependence.

The ac-dc difference of the transportable resistor was

determined by comparison with the coaxial 1000-0 resistor. The difference has changed from 484 X 10_9 in 1970

to 535 X 10-9 on January 29, 1974. In February or March

of 1Q74 the phase angle of the transportable resistor changed abruptly by about 43 /xrad. Subsequently, its ac-dc difference was determined to be 460 X 10-9

. The

dc resistance - of the transportable resistor was always

larger than the ac resistance. DIRECT CURRENT MEASUREMENTS

All of the dc comparisons relating the 1000-O transportable resistor to the NBS legal ohm as maintained with 1-0

resistors were made directly or under the supervision of T. E. Wells of the Electrical Reference Standards Section,

NBS Electricity Division. Two separate buildup processes were used. In the first process, a Hamon divider with ten

10-0 coils was first compared in the parallel mode with the

1-0 standard resistors using a current comparator. Then,

again using the current comparator, the Hamon divider was compared in the series mode with a second Hamon

divider containing ten 100-O coils, first to measure the

series-parallel combination of nine of the coils, and second

to measure the tenth coil individually. Finally, the second

Hamon divider was connected in the series mode and compared with the 1000-0 transportable resistor in a conventional Wheatstone bridge. In the second process, a Hamon divider with thirty-two 32-0 coils was compared

in the parallel mode with the 1-0 standard resistors using

a current comparator. Then it was compared in the series mode with a 1024-0 resistor consisting of a 100O-O resistor

in series with a 24-0 resistor. A separate measurement of the 24-0 resistor yielded the value of the 1000-0 section,

which was finally compared with the transportable resistor. The two processes were found to agree within the

uncertainty limits of the measurements, or about 2 parts

in 108

.

MEASUREMENT RESULTS

The measurement process consisted of performing the

ac calibrations relating the 10-pF transportable capacitor

to the 1000-O transportable resistor twice on a Monday or Tuesday, and twice again on the following Thursday ox Friday. On the intervening Wednesday, the 10-pF capici- tor was calibrated by reference to the calculable capacitor,

and the 1000-O resistor was compare with the bank of 1-0

resistors maintaining the NBS legal ohm. These sequences were repeated every month or two, with the intervals be- tween measurements being devoted to calibrating the

bridges and determining the required corrections as outlined earlier. Table I summarizes the results obtained. The standard

deviations associated with the average values of both

Fnbs and Onus are well below one part in 109

, which is negligible compared with the possible systematic errors. Estimates of the effects of possible systematic errors are itemized in Table II. The error in the measurement of Onbs is almost certainly less than the direct sum of the

listed systematic errors, but could easily exceed their root sum square (rss) sum. If a single number must be assigned

to the uncertainly of the measurement of Onbs, a value of 0.06 ppm (95-percent confidence level) would not be unreasonable. A comparison between the work reported here and the

absolute ohm work of the National Standards Laboratory

(NSL) of Australia [13] can be made by making use of the periodic resistance intercomparisons performed at the

International Bureau of Weights and Measures (BIPM).

The BIPM has reported for the series of intercomparisons

made between January and April 1973 that 0NBs =

069-bi + 0:2 M0, and 0NSL = 069 _Bi + 0.33 m0 [14]. Very

recent work at the NSL has indicated that the resistance

of the shield return lead of their ac-dc transfer resistor has increased by 0.19 ppm of the total resistance since it was last measured in 1967 [15]. This shield resistance was

included in their four-pair ac measurements, but its as- signed value was added to the results of their dc measurements. It is not known when the change occurred, but it is believed probable that most of the change occurred before

their measurement of the transportable resistors which

were involved in the BIPM resistance intercomparisons.

If the entire 0.19-ppm correction is applied to the 1973

intercomparisons, the BIPM result would become Onsl =

069-bi + 0.52 m0, so that Onbs = Onsl -0.32 juO. The

NSL unit is maintained to be in accordance with the SI

definition, but they had been using c = 2.997925 X 10s m/s for the speed of light. We have used c = 2.9979245S X

108 m/s, as was recommended for international adoption

in 1973 [16]. Since the speed of light enters the capacitance

equations as 1/c2

, the NBS unit of resistance as measured

through the BIPM by NSL should be corrected by 0.280 ppm, or Onus (NSL) = 10—0.60^0. The discrepancy

with our result, Onbs =10 —0.S19 fiQ, is 0.22 ppm. This is larger than expected in view of the reported NSL uncer147
cutkosky: nbs measurements TABLE I Summary of Measurements of Fnbs and Qnbs in SI Units,

Taking c = 2.99792458 X 108 m/s"

10-pF 1000-n Date Capacitor Fnbs Resistor ^NBS

9/12/73 +5.522 + 1.79S +9.409 -0.819

10/24/73 +5.542 + 1.798 +9.553 -0.822

11/28/73 +5.523 +1.775 +9.C67 -0.821

1/23/74 +5.533 +1.784 +9.832 -0.708b 4/3/74 +5.532 + 1.782 +10.073 -0.813

average: +1.787 average of four: -0.819

a Corrections in parts per million to indicated quantities or stand- ards are tabulated.

b This measurement was discarded because the dc comparison

exhibited internal discrepancies as large as 2 parts in 107.

TABLE II Estimates of the Effects of Possible Systematic Errors on

the Measurement of Qnbs (95-Percent Confidence Level)

Error Estimates

Possible Systematic Errors (ppm)

Drift in laser wavelength between calibrations 0.010

Imperfection of calculable capacitor electrode straightness and alignment 0.010 Misalignment of laser beam in capacitor 0.002

Diffraction error due to laser aperture diameter 0.001 Load corrections for calculable capacitor 0.007

Residual gas in calculable capacitor 0.004

Drift between calibrations and measurement uncertainties

of transformer ratios 0.007

Uncertainty in capacitance voltage dependence measure- ment ' 0.003 Temperature uncertainty of transportable resistor 0.004 Frequency dependence of transportable resistor 0.005

Self-heating effects in transportable resistor 0.004

Uncertainty in dc stepup, 1 to 1000 ft 0.050

direct sum: 0. 107

rss sum : 0 . 054

tainty of 0.2 ppm and our own estimated uncertainty of 0.06 ppm, and in view of the fact that a direct intercomparison of capacitance standards between NBS and NSL

indicates agreement within 0.02 ppm. This implies that very little of the discrepancy can be attributed to the cal- culable capacitors.

A second intercomparison of the NBS unit of resistance with that of the BIPM was made in May 197-3 and indi- cated that Onus = ^69-bi —0.17 p£l [17]. This result was

not used in the formal assignment of the NBS unit of re- sistance in terms of the BIPM unit because the first inter- comparison seemed to be in closer agreement with the trend through measurements dating back to 19~>7, and

because some comparisons of 10M> resistors also indicated

closer agreement with the first series of intercomparisons.

If the second intercomparison is taken to be correct, one would obtain, using the same corrections as above, Onus (NSL) = lft-0.97 )xP.. which differs from the NBS result by 0.15 ppm, but with the opposite sign.

It is clear that further study of the resistance transfer problem must be made. Although the precise cause of the discrepancy is not yet known, it has been noticed that the NBS transportable resistors do not return immediately

to their original rates after their return from other laboratories. It has been suggested that the problem may be re- lated to the fact that NBS normally measures its resistors at 25°C and determines their values at 20°C through a

fairly quick series of temperature coefficient measurements,

whereas the BIPM normally measures the resistors after a rather long stabilization period at 20°C. This difference

in technique cannot explain all of the discrepant results, but points to the resistance transfer as the least reliable part of the intercomparison process. A direct interchange

of resistance standards between NBS and NSL which is now underway may help resolve the discrepancies.

REFERENCES

(1) A. M. Thompson and D. G. Lampard, "A new theorem in elec- trostatics and its application to calculable standards of capacitance," Nature, vol. 177, p. 888, 1956.

[2] R. D. Cutkosky, "Evaluation of the NBS unit of resistance based on a computable capacitor," NBS J. Res., vol. 65A, pp.

147-158,^ May-June 1961.

[3] , "Techniques for comparing four-terminal-pair admittance

standards," NBS J. Res., vol. 74C, pp. 63-78, July and Dec.

1970.

[4] , "Four-terminal-pair networks as precision admittance and impedance standards," Conunun. Elec, vol. 70, pp. 19-22,

Jan. 1964.

[5] R. D. Cutkosky and L. H. Lee, "Improved ten-picofarad fused

silica dielectric capacitor," NBS J. Res., vol. 69C, pp. 173-179,

July-Sept. 1965.

[6] R. D. Cutkosky, "A varactor null detector for audio frequency

capacitance bridges," IEEE Trans. Instrum. Meas., vol. IM-17,

pp. 232-238, Dec. 1968.

[7] , "An ac resistance thermometer bridge," NBS J. Res.,

vol. 74C, pp. 15-18, Jan. and June 1970.

[8] W. K. Clothier, "A calculable standard of capacitance,"

Metrologia, vol. 1, pp. 36-55, Apr. 1965.

[9] R. D. Cutkosky and J. Q. Shields, "The precision measurement

of transformer ratios," IRE Trans. Instrum., vol. 1-9, pp. 243-

250, Sept. 1960.

[10] J. Q. Shields, "Voltage dependence of precision air capacitors," NBS J. Res., vol. 69C, pp. 265-274, Oct.-Dec. 1965.

[Ill R- J- Haddad, "A resistor calculable from dc to u> = 105 rad/s," M.S. thesis, School Eng. Appl. Sci., George Washington Univ.,

Apr., 1969.

[12] C. G. M. Kirbyand M. .1. Laubitz, "The error due to the Peltier

effect in direct-current measurements of resistance," Metrologia,

vol. 9, pp. 103-106, Sept. 1973.

[13] A. M. Thompson, "An absolute determination of resistance based on a calculable standard of capacitance," Metrologia,

vol. -1, pp. 1-7, Jan. 1908.

[14] G. Leclerc, Rapport sur la 13' comparison ties etalons nationaux

de resistance electrii/ue, Bur. Int. Poids Mesures, Sevres, France, Jan.-Apr. 1973.

[15] Private communication, June 7, 1974.

[16] J. Terrien, " International agreement on the value of the velocity

of light," Metrologia, vol. 10, no. 1, p. 9, 1974.

[17] Private communication, Nov. 21, 1973.

148
ELECTRICAL STANDARDS

From over the world did they roam, Trading thoughts contained in the dome. The Ampere and Volt Were given a jolt; Then most of them quickly went ohm. Anon.

The Fundamental Electrical Standards: Present Status and Prospects

for Improvement

Robert D. Cutkosky

Institute for Basic Standards, National Bureau of Standards, Washington, D. C. 20234

Absolute measurements of resistance and current have until recently been based on instruments

involving carefully constructed and measured single layer solenoids. Minor improvements in tech- nique have resulted in a continuous reduction of measurement uncertainty, but overall uncertainties much smaller than one part in 10 6 are not likely to be achieved without radical changes in design.

Some of the principal sources of uncertainty in these experiments are analyzed, and, when possible, alternative measurement systems are proposed which could lead to higher accuracy. Within the time

scale appropriate for measurements of this kind, absolute resistance measurements based on

calculable capacitors are quite recent, even though the first such measurement was reported in 1961. More recent calculable capacitors have resulted in absolute resistance measurements of higher accuracy than may reasonably be expected from systems utilizing calculable solenoids. The

possibility for improvements in absolute current or absolute voltage measurements does not seem

quite so clear, but some ideas are presented which are either now being tried or are worth some consideration. An attempt is made to indicate the more important sources of uncertainty in the proposed

systems. Key words: Capacitors; current balances; electrical measurements; electrical standards; electrometers; inductors.

149
1. Introduction

It is unfortunate that the completion of experimental determinations of physical quantities and of

measurements related to them do not always coincide

with the convening of an international conference of

appropriate stature. In the case of electrical standards, many years commonly elapse between the

inception of a particular piece of work and its completion, and in view of the fact that only a few

laboratories in the world are involved in such measurements, it may not be surprising that this paper

contains no results. In the short space available I shall quickly summarize the more recent absolute measurements of

capacitance, resistance, voltage and current, and

give the most attention to the present status of

work now underway and to various proposals for improved measurement systems. Some comments on

the possible future use of Josephson junctions as maintenance standards of voltage will also be made,

although physically the Josephson effect is most

useful as a means of accurately measuring e/h.

The starting point for all absolute electrical measurements rests in the extensive body of classical electromagnetic theory, by means of which one can

in principle calculate the magnetic fields, inductances,

and forces associated with current-carrying conductors, and the electric fields, capacitances, and

forces associated with charged conductors, from

mechanical measurements of the shapes and positions

of these conductors. The task of the metrologist

involved in this work is to determine a suitable

geometry for his equipment so that the necessary

mechanical measurements can be made with a high

accuracy, and so that the resultant electrical property has such a magnitude that it can be

accurately compared with a set of fixed "maintenance" standards which preserve the unit so obtained. Many compromises must be made in the

design of absolute electrical measurement equipment to minimize the uncertainty of the final result. These compromises involve among other things

considerations of the relative ease with which

various systems can be fabricated and measured.

2. Units of Impedance

Two quite distinct approaches to an absolute ohm

measurement are available. One approach begins

with an inductor whose inductance can be calculated

from its mechanical dimensions, and the other begins

with a capacitor whose capacitance can be calculated

from its mechanical dimensions. The second method

requires a knowledge of the speed of light, which is generally treated as an auxiliary constant. 1 In

principle either a calculable inductor or a calculable

capacitor can be used to assign a value to a standard

resistor using purely bridge methods, and requiring

only a measurement of frequency in addition to

1 The vacuum capacitance of a capacitor is equal to a geometric factor times the electric constant €o, where €o/io=l/c2

. The magnetic constant /xo=47rX10-7 henry /meter in the SI system.

suitable impedance ratio measurement techniques.

Generally these impedance comparisons can be made

with substantially better accuracy than is presently

achievable in the construction and evaluation of

either a calculable inductor or a calculable capacitor.

2.1. The Ohm from Inductance

Calculable self inductors are generally constructed

by preparing a grooved cylindrical former of a stable material such as fused silica, and placing a singlelayer helical winding of copper wire in the groove. The dimensions of the completed solenoid are measured as precisely as possible and from these

measurements the self inductance of the solenoid can be calculated. Calculable mutual inductors usually contain one

precision solenoid similar to that described above,

but with one or more gaps in the helical winding,

placed so that a region exists exterior to the solenoid

in which the magnetic field is very small. The

secondary winding is placed in this low-field region,

since it is then not necessary for its dimensions

or its location to be measured with a high accuracy. Multilayer windings can be used for the secondary

without seriously affecting the accuracy with which

the mutual inductance can be calculated. Alter- natively, a mutual inductor can be constructed by

placing a bifilar winding on a single cylindrical

former.

At least six absolute ohm determinations based on

calculable inductors were reported since 1955. Most

of these measurements utilized mutual inductors

rather than self inductors, partly because self inductors have larger phase angles than mutual

inductors, which complicates the electrical part of

the measurement. Two of the recent inductor-based

absolute ohm determinations were assigned uncertainties of about 2 parts in 106 [1, 2]. In both

cases, the uncertainties were almost entirely at- tributable to the difficulties involved in measuring

the mechanical dimensions of the single-layer, wire

wound solenoids.

Some reduction in the mechanical measurement

uncertainties for solenoids may be possible, but it seems likely that an uncertainty substantially

smaller than 1 part in 106 will not be achievable

with present inductor designs. The reasons for this are first, that even single layer solenoids wound on

rigid, grooved formers are slightly compressible,

and uncertainties remain even after corrections are applied for the distortion caused by the pressure

of the micrometer used in the measurement of

solenoid diameter and pitch. Second, oxide films on

the wire cause formidable problems; and third, the

distortion of the wire caused by winding it tightly on the former cannot be easily determined. Finally,

it is necessary to estimate the current distribution

in the wire. This depends upon how the resistivity

of the wire varies with distance from the axis of the

solenoid, and is a function of wire diameter, solenoid

diameter, winding tension, and the material chosen

for the wire, which is usually copper. Eddy currents within the wires also affect the current distribution,

150
and are not easy to estimate (eddy currents induced

in other metal parts of the system can also cause

serious errors if the problem is not recognized).

The current distribution uncertainties discussed

above could probably be reduced by reducing the

wire diameter; but if carried to an extreme, the wire

distortion caused by micrometer pressure would be

very large. Additionally, reducing the wire cross section would increase the solenoid resistance, and

result in either an increase in power consumption

and hence larger temperature uncertainties if the

solenoid is in a current carrying part of the electrical

circuit, or in increased thermal agitation noise if it

is in the potential part of the circuit. Proposals have been made to produce solenoids by

evaporating very thin metal films in a helical pattern on smooth cylindrical forms. There is no doubt that

this would result in a reduction of the solenoid

diameter uncertainty, but it would be very difficult to determine how the current was distributed over

the width of the film. Very small defects in the

film could cause significant local variations in the

pitch of the solenoid, which could result in substantial errors in the calculated inductance. The

relatively large solenoid resistance to be expected

with such a system could also be a problem for the reasons cited above. It has been argued that the

use of superconducting windings on a solenoid would

eliminate many of the above problems, but the cur- rent distribution within a superconducting wire is at best ambiguous, and no reasonable way has yet been

found for measuring the solenoid dimensions at low temperature.

A completely different geometry for a calculable

mutual inductor has been treated theoretically by

Page [3]. The proposed system uses a set of eight

parallel wires equally spaced around a circle. In this system, the mutual inductances between each pair

of wires and every other pair of wires is measured,

and these mutual inductances can be related to the length of the wire system. Many of the uncertainties due to small errors in wire placement are of

second order, and only the length of the system need

be measured to high accuracy. No serious attempt

has yet been made to construct such a system, but it seems clear that the problems would be formidable.

The mutual inductances are likely to be in the order

of 10-7 H, which presents serious but not insurmountable electrical measurement problems E4J.

2.2. The Ohm from Capacitance

Calculable capacitors have three principal advantages over calculable inductors. Negligible power

is lost in a capacitor, so the mechanical dimensions

do not change with time due to the resultant change

in temperature. The corresponding problem with

capacitors has to do with dimensional changes due

to the electrostatic forces between the capacitor

electrodes, and usually contributes errors smaller

than one part in 108

. The second advantage of

capacitors is that the current distribution problem

does not exist. In this respect, capacitors are not as pure as might be wished, because dielectric films on

the electrode surfaces introduce an analogous

uncertainty in the effective electrode spacing.

However, the dielectric films are usually quite thin,

and the resultant uncertainties are much smaller

than the uncertainties resulting from the current

distribution in a calculable inductor. The third

advantage of calculable capacitors over calculable

inductors is that all critical fields are contained

within the calculable region, so that the effects of

objects external to the calculable instrument can be

ignored. This eliminates the need for special non- magnetic buildings as are generally required for highprecision inductance measurements.

All calculable capacitors now in existence or known to be under construction are of a type called a cross capacitor. This geometry was first proposed

by A. M. Thompson and D. G. Lampard, and in practice is usually constructed of four cylindrical

electrodes of circular cross section [5j. The axes of

the four electrodes are parallel and are located at the

corners of a square just slightly larger than the

electrode diameter. The two capacitances of interest are the capacitance coefficients between opposite

pairs of electrodes. The exterior of the system is completely shielded so that only the region between

the four electrodes is of concern. Provided that reasonable care is taken to obtain

the requisite two-dimensional symmetry, all small

perturbations from the ideal geometry including

those due to dielectric films on the electrode surfaces enter only to second order in the average of the two

cross capacitances, leaving only the first order

dependence of capacitance upon the overall length of

the electrode system. This single critical dimension

is measured in the more refined cross capacitors

with interferometric techniques.

Several cross capacitors have been described in recent years, but so far only two absolute ohm

measurements based on cross capacitors have been

completed [6, 7~]. The most accurate of these two

was the ohm measurement reported by Thompson

and making use of the cross capacitor described by

Clothier [8]. They reported an overall uncertainty

of one part in 107 exclusive of the uncertainty in the

speed of light. Somewhat smaller uncertainties may

be expected soon as improvements are made in the

Clothier/Thompson apparatus and as similar systems now under construction elsewhere are completed.

At the present time it appears that capacitor-based

absolute ohm measurements are capable of pro- viding much higher accuracy than are inductor-based

absolute ohm measurements. The relative accuracies

of the two techniques are so disparate that work with

calculable inductors could well be abandoned, except

that a comparison of measurements made in such

completely different ways helps to prevent gross

errors. Inasmuch as a capacitor-based absolute ohm

measurement requires a knowledge of the speed of

light and an inductor-based measurement does not, one can treat the pair of ohm measurements as a speed -of-light measurement, and possibly learn

something about the dependence of c upon frequency.

No discrepancy attributable to such an effect has

151
yet been observed, and the continuation of calculable

inductor research on this ground might be hard to

justify.

3. Units of Voltage or Current

After having established a unit of resistance by means of one of the techniques outlined above, an

absolute determination of either the volt or the

ampere suffices to establish both units through

Ohm's law. In practice it is the unit of voltage that is maintained rather than the unit of current, because

of the proven stability of saturated cadmium

sulfate standard cells, but in fact all of the precision

measurements of this kind made until now have been

with current sensitive instruments called current

balances. Voltage can also be measured directly with

electrometer-type instruments, and some such instruments are now under construction.

From a dimensional standpoint, absolute resistance

measurements require only length and time deter- minations, whereas voltage or current measurements

require in addition a determination of mass. The

experiments involve a measurement of the torque

or force between two current carrying conductors

or between two charged electrodes. The electrical force is usually compared with the gravitational

force on a known mass, and hence a knowledge of the

local acceleration of gravity is also required. These

additional complications tend to make absolute

voltage or current measurements much more difficult than absolute resistance measurements.

3.1. Current Balances

The most common type of current balance is often called a Rayleigh balance, and contains a

fixed center-tapped solenoid having equal currents

in each half of the solenoid but of opposite direction, to yield a strong radial component of magnetic

field in the center of the system. A relatively short

current-carrying solenoid is placed in the center of

the system and attached to one arm of a precision

balance. Reversing the sign of the fixed-solenoid current while keeping the current in the movable

solenoid fixed changes the sign of the force on the

movable coil. The balance is then restored to equilibrium by adding a known mass to that arm of

the balance. Some Rayleigh balances have a set of

coils on each arm of the balance, and the movable

solenoid (or solenoids) may be either of larger or smaller diameter than the fixed solenoid. Within the

last few years, about four Rayleigh-type current balances have been described, with the smallest

reported uncertainties being about 4 parts in 106 [9], and about 5 parts in 106 [10].

One current balance has been described which

involves the torque between two current carrying

solenoids whose axes are perpendicular [11]. The

torque is measured by attaching one solenoid to a balance beam and restoring the balance equilibrium

after reversal of the current in the fixed solenoid by

placing a known mass on one pan of the balance.

The distance between the knife edge supporting this

pan and the central knife edge of the balance must

also be measured with this system. Some problems

were encountered initially with distortion of the

balance beam, but this probably does not constitute

a fundamental limitation of the method. The most

recent ampere determination with this instrument

yielded an uncertainty of 7 parts in 106 [12].

The solenoids for all types of current balance are normally constructed and measured using the same

techniques as are used for calculable inductors,

and all of the problems encountered in calculable

inductors also occur in current balances, except that

since direct current is used in current balances, eddy

current effects do not influence the current distribution in the wires. Since the movable solenoid of a

current balance must be kept as light as possible to avoid overloading the balance, the movable solenoid

is usually small and quite compressible, and its dimensions are therefore difficult to measure ac- curately. This is a principal contributor to the

uncertainty in a current balance experiment.

A second major problem encountered in current balances is caused by air currents generated by the

heated wires, which disturb the balance. Presumably no systematic errors are caused by the air currents,

but very erratic balance fluctuations are observed,

and long measurement sequences are required in order to obtain a reasonably small uncertainty in the

measured force or torque.

Until recently the uncertainty in the local accelera- tion of gravity contributed substantially to the

overall uncertainty of an ampere determination.

Recent refinements in gravity measurement tech- niques have made this source of uncertainty negligible.

A substantial reduction in the uncertainty of an

absolute ampere determination would seem to

require an improved technique for constructing and

measuring the movable solenoid. The fixed solenoid can be made large and rigid enough to allow a rela- tively accurate determination of its dimensions, and

schemes exist for comparing the pertinent electrical properties of one solenoid with another. For example,

the ratio of the magnetic fields in the centers of

two solenoids carrying equal currents is a function

of the relative dimensions of the two solenoids, and

may be used in conjunction with a set of mechanical

measurements of both solenoids to improve the data

obtained for the less precisely known solenoid by a

least squares process. Only a small improvement in the calculated force constant of a current balance is likely to result unless the two solenoids are specifically designed to take advantage of this tech- nique.

Another approach to a better ampere determination involves placing the movable solenoid in a much stronger field than can be conveniently obtained from a calculable fixed solenoid, and measuring the ratio of this strong field to the much weaker

field of a calculable solenoid using proton gyro- magnetic resonance techniques. Forces strong enough

to be accurately measured can be obtained with this system using a much simpler and more precisely

152
measurable movable solenoid, and in fact, the

movable "solenoid" usually consists of a winding

around the edge of a rectangular insulating form. In this case an evaporated metal "winding" might

offer some advantages, since the distribution of current over the width of the conductor is of little concern. A possible flaw in the above system (sometimes

called a Cotton balance) is that one must assume

that the proton gyromagnetic ratio is independent

of field strength. No contrary evidence exists, but

lacking further verification, the results of such an

experiment might be subject to question. It is interesting to note that the most direct evidence for the independence of the proton gyromagnetic ratio

upon field strength is based on measurements

conceptually similar to those described above and on measurements with a conventional current balance.

Measurements by Huggins and Sanders of the ratio

of the proton resonance frequency to the resonance

frequencies of 3 H, 7 Li, nB, and 19 F at both high and

low field intensity indicated that the field dependence

of the proton resonance frequency is certainly small,

but a slight field dependence cannot be ruled out [13].

3.2. Electrometers

The possibility of obtaining the unit of voltage by

measuring the force between two charged electrodes

has long been intriguing, because of the absence of the

air currents which disturb a current balance. Unfortunately the electrometer geometry appropriate

for such a system is not usually amenable to the

precise mechanical measurements necessary for determining the force constant with adequate

precision directly, and this fact has discouraged

work in this direction. One proposed scheme for constructing a directly calculable electrometer utilizes a pool of mercury

with a fixed flat horizontal electrode just above the

pool [14]. The distance between the electrode and

the pool can be measured interferometrically with

and without a voltage between the pool and the

electrode. The electrical force between the pool and

the electrode can then be inferred from measurements

of the density of the mercury and of the local acceleration of gravity.

Problems are expected with this system in measuring the mercury density to an adequate accuracy, and in assuring the cleanliness of and lack of surface

films on the mercury. The system appears to be

feasible, and preliminary investigations have indi- cated that the mercury surface can be maintained

reasonably stable and ripple-free, but the overall uncertainty likely to result cannot yet be estimated.

An indirect approach to an electrometer system

makes use of electrodes in the form of interleaving

cylinders, for which the electrical force is constant

over a substantial axial displacement of one electrode

with respect to the other. The force at a given

electrode spacing is proportional to the rate of

change of capacitance with respect to distance,

and since the force is nearly constant, one can show

that the average force over a finite electrode dis- placement is equal to ^V2 AC/ A/. It is proposed that

the total electrode displacement A/ be measured

interferometrically, and that AC be measured by

comparison with a calculable capacitor. The average

force can presumably be determined from a large

number of discrete force measurements made with

different relative electrode displacements.

The above system seems relatively straightforward at first glance, but some difficult problems

remain unsolved. For example, if the voltage between the electrode is dc, the capacitance required

for use in the equations is the d-c value, and may

differ from the more readily measurable ac capaci- tance obtained in terms of a calculable capacitor

using conventional bridge techniques. Slightly lossy

dielectric films on the surfaces of the electrodes

would produce such a difference, and hence great care would be necessary to obtain clean electrodes. This problem would be avoided if the electrodes were excited with alternating current, but the

substantial double frequency force variation that

would then result might prove damaging to the knife

edges of the balance, and in addition the measured ac voltage would have to be transferred to dc with a thermal transfer instrument for comparison with the

voltage of a standard cell. Any electrometer would be improved if it could be

operated in a vacuum, because the buoyancy cor- rections for the mass and the dielectric constant

correction for air would both be eliminated. The

mercury-pool electrometer would probably work well under these conditions, although evaporation of the

mercury has been reported to cause some problems.

Experiments involving the operation of conventional

knife-edge balances in a vacuum unfortunately have

not been notably successful. It is expected that

occluded surface films on the mass standard would be

partially released under vacuum conditions, making

an assignment of the vacuum mass a rather difficult task. The magnitude of this effect is at the present

time not known.

In view of the many difficult problems that one can

foresee in the construction and operation of a precision electrometer, it is not possible at this time

to estimate the accuracy likely to be obtained.

Judging from some preliminary investigations, it seems to be possible to obtain the unit of voltage

from an electrometer with an uncertainty at least as small as can now be obtained by means of a current balance, or a few parts in 106

.

4. Standards for Maintaining Electrical

Quantities

There is little point in investing a great deal of

effort in an absolute electrical measurement if there

are no adequate standards for maintaining the unit, or for use in comparing the measurements made in different laboratories. The currently accepted standards for maintaining the units of voltage and

153
resistance appear to introduce uncertainties of a few

parts in 107

. Some possibilities exist for improving

these standards, but at present only minor improvements can be foreseen.

4.1. Maintenance Standards for Impedance

The unit of resistance has been maintained by means of 1-ft, dc resistors for at least fifty years. Proposals have been made to use 104 -ft resistors for

this purpose because of the valid argument that they

are not as strongly influenced by thermal voltages

in the measurement circuit, and because of the

dubious argument that the optimum source impedances of some recently developed galvanometers

are closer to 104 ft than to 1 ft. Improvements in lowimpedance galvanometers in the next few years

cannot be ruled out, but the uncertainties caused by

varying thermal voltages will always be more

significant with smaller resistors. If careful attention

is given to the construction of 1-ft resistors and the

associated measurement circuit, one might expect

to reduce the uncertainties due to thermal voltages

to about one part in 108

, whereas the limit set by

thermal agitation noise is closer to one part in 109

, and is independent of the resistance. Since a cal- culable capacitor is inherently a high impedance

standard, it is somewhat easier to measure a 104 -ft resistor in terms of such a standard than it is to measure a 1-ft resistor. In practice the transfer from

ac impedance to dc impedance is usually made at the 1000-ft level, and as a result only one less measure- ment is required if the unit is maintained at 104

ft. A procedure for accurately measuring 1-ft resistors is required in any case, since current balances normally

employ currents of about 1 A, which are measured

by comparing the voltage produced by this current

passing through a 1-ft resistor with the voltage of a standard cell. The effects of thermal voltages could be eliminated

by maintaining the unit of resistance by means of ac

resistors. Existing ac detectors and matching net- works are sufficiently well advanced to allow the

measurement of any ac resistor between 1 ft and 105 ft with an uncertainty limited only by thermal agitation noise in the resistor. Capacitors can also be

used to maintain a unit of ac impedance, and

measurement uncertainties of about one part in 109 can in this case be achieved. The absence of self- heating effects in capacitors makes them especially

appealing. Existing capacitors appear to drift less than one part in 107 per year, but are not quite as stable as the best 1-ft resistors. It is probably too early to recommend a change in the mode of maintaining units of impedance. The

most practical solution is for laboratories to maintain

separate sets of promising standards for several

years, and to periodically intercompare the sets with each other. If a particular type of standard is then found to be more suitable than the others, this fact will emerge from a statistical analysis of the

data, and no formal abandonment of any type of

standard need be made. The maintained unit could

be a weighted average of the units maintained by the

different types of standards, with the weighting

factors changing with time as stability data ac- cumulates. If the uncertainties associated with inter- comparing the different sets of standards were very

much larger than the uncertainties associated with

comparing the standards within a set with each

other, it could be useful to recognize more than one maintained unit for a single quantity, but in this case it would be very important to distinguish

between the various units.

4.2. Maintenance Standards for Voltage

The unit of voltage is universally maintained by means of saturated Weston cells. These standards

are much more stable than any other known electro- chemical systems or semiconductor devices. The

voltages of saturated Weston cells unfortunately

return to equilibrium very slowly after a thermal

disturbance, so that great care is needed when they

are transported between laboratories. Because of the

large temperature coefficients of voltage characteristics of these cells, and especially because of the

very large differential temperature coefficient between the two legs of a cell, very careful temperature

control is required.

Recently it has become common to transfer the

unit of voltage between laboratories by means of

groups of standard cells contained in temperatureregulated enclosures which are maintained at constant temperature during shipment. A significant

reduction in the uncertainty assigned to a transferred

unit of voltage can be made by means of these

systems. Unfortunately, with present enclosures the

cell temperatures are not maintained as precisely as could be desired.

A great surge of activity is now taking place in many laboratories to determine whether or not the

voltage steps induced by microwave radiation in the

d-c current of a Josephson junction can be used to maintain the unit of voltage. Present indications

are that Josephson junctions may well prove to be

superior to saturated Weston cells for this purpose. An added attraction for this technique is that the

use of Josephson junction voltage references would

eliminate the need for transporting the unit of

voltage between laboratories, since in principle only a published value for e/h and a reference to a time

standard are necessary to determine the voltage at

a given step of a Josephson junction. The major

experimental problem encountered with Josephson

junctions is that of accurately comparing their rather small voltages (presently of the order of

10~2 V) with the voltages of standard cells. The

ability to make this comparison with high accuracy

is essential in order to tie the voltage unit maintained with Josephson junctions to the absolute

voltage unit obtained by means of a current balance.2

2 The alternative possibility of defining the value of e/h would not eliminate

the necessity for making measurements with current balances or electrometers, and would in fact degrade the accuracy of the ohm [15]

154
5. References

[1] Rayner, G. H., An absolute Determination of Resistance by Campbell's Method, Metrologia 3, No. 1, 12-18

(Jan. 1967).

[2] Linckh, H. E., und Brasock, F., Eine Methode Zur Bestimmung des Widerstandswertes aus der Induktivitat, Metrologia 4, No. 3, 94-101 (July 1968).

[3] Page, C. H., A New Type of Computable Inductor, J. Res. Nat. Bur. Stand. (U.S.), 67B, No. 1, 31-39 (Jan-Mar

1963).

[4] Homan, D. N., Some Techniques for Measuring Small

Inductances, J. Res. Nat. Bur. Stand. (U.S.), 70C, No. 4, 221-226 (Oct.-Dec. 1966).

[5] Thompson, A. M., and Lampard, D. G., A New Theorem

in Electrostatics and its Application to Calculable Standards of Capacitance, Nature 177, 888 (1956).

[6] Cutkosky, R. D., Evaluation of the NBS Unit of Resist- ance Based on a Computable Capacitor, J. Res. Nat.

Bur. Stand. (U.S.), 65A, No. 3, 147-158 (May-June

1961).

[7] Thompson, A. M., An Absolute Determination of

Resistance Based on a Calculable Standard of Capacitance, Metrologia 4, No. 1, 1-7 (Jan. 1968).

[8] Clothier, W. K., A Calculable Standard of Capacitance,

Metrologia 1, No. 2, 36-55 (April 1965).

[9] Vigoureux, P., A Determination of the Ampere, Metrologia 1, No. 1, 3-7 (Jan. 1965).

[10] Driscoll, R. L., and Cutkosky, R. D., Measurement of Current with the National Bureau of Standards

Current Balance, J. Res. Nat. Bur. Stand. (U.S.), 60, No. 4, 297-305 (April 1958).

[11] Driscoll, R. L., Measurement of Current with a Pellat- Type Electrodvnamometer, J. Res. Nat. Bur. Stand.

(U.S.), 60, No. 4, 287-296 (April 1958).

[12] Driscoll, R. L., and Olsen, P. T., Unpublished Communication to the Comite Consultatif D'EIectricit6.

[13] Huggins, R. W., and Sanders, J. H., Nuclear Magnetic

Moment Ratios Measured in High and Low Fields, Proc. Phys. Soc. 86, 53-63 (1965).

[14] Clothier, W. K., A Proposal for an Absolute Liquid

Electrometer, Metrologia 1, No. 4, 181-184 (Oct. 1965).

[15] Page, C. H., Letters to the Editor, Metrologia 1, No. 2, 73-74 (April 1965). DISCUSSION

U. Stille: May I just mention one point which

touches the general title of our conference, fundamental constants. You mentioned regarding the reali- zation of the ohm differing methods using inductance

and using capacitance. There is in effect a difference

between these two methods. If you are thinking

about accuracies higher than 10-8 or in the region of

10-8

, then there is a special need. You see, inductance

in our system of describing physics to which the SI

units are appropriate enters only the magnetic constant which by the definition of the ampere indirectly

got a fixed value. But regarding capacitance there

enters the electric constant or, in other words,

the square of the velocity of light. So then you

must know the value of the velocity of light to this accuracy. R. D. Cutkosky: That is correct. Comment from the floor: Measure it with fre- quency. (Laughter)

155

ABSOLUTE VALUES OF ELECTRICAL UNITS - II

Realization of Electrical Units: Quantum Methods

A Test of the Quantum Hall Effect as a Resistance

Standard

Marvin E. Cage, Ronald F. Dziuba and

Bruce F. Field (1984) 159

A High Accuracy Automated Resistance Bridge for

Measuring Quantum Hall Devices

Bruce F. Field (1984) 162

A Possible Quantum Hall Effect Resistance Standard

M. E. Cage, R. J. Dziuba and B. F. Field (1 983) 165

The Quantum Hall Effect, I M. E. Cage and S. M. Girvin (1 983) 170

The Quantum Hall Effect, II S. M. Girvin and M. E. Cage (1 983) 185

Progress Report on Investigations of the Quantum

Hall Effect as a Possible Resistance Standard

M. E. Cage, et al. (1 983) 197

Laboratory Voltage Standard Based on 2e/h

Bruce Field and Victor W. Hesterman (1976) 208

Cryogenic Voltage Comparator System for

2e/h Measurements

Ronald F. Dziuba, Bruce F. Field and

Thomas E. Finnegan (1974) 211

Volt Maintenance at NBS via 2e/h: A New Definition

of the NBS Volt

B. F. Field, T. F. Finnegan and J. Toots (1 973) 215

Summary of International Comparisons of

As-Maintained Units of Voltage and Units of 2e/h

Woodward G. Eicke, Jr. and Barry N. Taylor (1972) 227

A Flexible System with Two Selectable Ratios for

Use with Josephson Devices

David W. Braudaway (Abstract, 1978) 231

High Accuracy Potentiometers for Use with Ten

Millivolt Josephson Devices: II - Cascade

Interchange Comparator

A. Denenstein and T. F. Finnegan (Abstract, 1974) 231

157
High Accuracy Potentiometers for Use with Ten

Millivolt Josephson Devices: I - Double SeriesParallel Exchange Comparator

T. F. Finnegan and A. Denenstein (Abstract, 1973) • •

AC Josephson Effect Determination of e/h:

A Standard of Electro-Chemical Potential Based on Macroscopic Quantum Phase Coherence in Superconductors

T. F. Finnegan, A. Denenstein and

D. N. Langenberg (Abstract, 1971)

158
A Test of the Quantum Hall Effect as a

Resistance Standard

MARVIN E. CAGE, RONALD F. DZIUBA, member, ieee, and BRUCE F. FIELD, member, ieee

Abstract -This paper demonstrates that the quantum Hall effect can

be used to monitor a laboratory unit of resistance. A 6453. 2-n room

temperature reference resistor was calibrated relative to two quantum

Hall effect devices with a 0.017-ppm (la) uncertainty for each 1 h measurement period. This accuracy was achieved by correcting for a measurement system offset error and for the temperature dependences

of each quantum Hall device. Hamon series-parallel resistor networks

were then used to calibrate the 6453. 2-n resistor in terms of the five one ohm resistors which comprise the NBS ohm. The total la accuracy

for the transfer between the quantum Hall devices and the 1-12 resistors was 0.047 ppm.

Manuscript received August 8, 1984. This paper was supported in part by the U.S. Office of Naval Research. The authors are with the Electricity Division, National Bureau of

Standards, Gaithersburg, MD 20899.

T

[1]:

I. Introduction

HE HALL resistance RH of a two-dimensional electron

gas is, under certain conditions, quantized in units of h/e2

Rh(!) =

v. h 25812.80

e

2

i (1)

where VH is the Hall voltage across the sample, / is the current through the sample, h is the Planck constant, e the elementary

charge, and i is an integer quantum number. Ultimately one would like to determine the value of RH in order to verify that

it is indeed equal to submultiples of h/e 2 [2] and thus depends

only on fundamental constants of nature. If this were proved

U.S. Government work not protected by U.S. Copyright

159
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

250 | 1 r

200

~ I 50

100

3 4 5 MAGNETIC FIELD (T)

Fig. 1. Recording of Vyj and Vx versus magnetic field for a GaAs

device cooled to 1.2 K. The current is 25.5 nA.

to be correct, then the quantum Hall effect could be used as an absolute resistance standard. An an intermediate step in achieving this goal it is necessary

to demonstrate that the quantum Hall effect can be used as a

relative standard to maintain a laboratory unit of resistance based on wire-wound resistors analogous to the way in which

the ac Josephson effect is used to maintain a laboratory unit of voltage. To be of practical use this standard would need to be capable of calibrating resistors to a relative accuracy of a few parts in 10 8

. We demonstrate that this requirement is indeed achievable.

II. Quantum Hall Effect Measurements

Two high-quality, quantum Hall effect devices were used; both were GaAs-Alx Gaj _ x As heterostructures grown by molecular beam epitaxy by A. C. Gossard at AT&T Bell Laboratories, and then prepared into Hall bars and screened by D. C. Tsui at Princeton University, Princeton, NJ. They are 4.6 mm

long and 0.38 mm wide, and have three sets of potential probes, with two sets symmetrically displaced ± 1 .0 mm along the channel from the center set. Their zero magnetic field mobilities were ~ 10 s cm2 /(V • s) at 4.2 K. Both devices were optimized

for the i = 4 Hall step, where RH(4) « 6453.2 ft. Fig. 1 shows

a low-sensitivity recording of VH and Vx versus magnetic field for one of the devices for a current of 25.5 jiA at 1.2 K. The

VH and Vxplots were equally as good for the second device. The first part of the experiment consisted of making potentiometric comparisons of the quantum Hall voltage VH with

the voltage drop VR across a series-connected room temperature reference resistor using the measurement circuit indicated

in Fig. 2. The wire-wound reference resistors were adjusted to have a resistance within a few parts-per-million (ppm) of RH. They were then hermetically sealed in silicone-fluid-filled containers and placed in temperature regulated enclosures controlled to within ±0.002°C of a nominal temperature of ap- proximately 28°C. To measure Rfj in terms of a reference resistor, the poten- tiometer voltage is made almost equal to the voltage drop

across VH or VR . An electronic detector, D, with an input

CONSTANT CURRENT

SOURCE

REF. RESISTOR ; 6.453.20 SI

-® 1

— POTENTIOMETER

J

Fig. 2. A simplified schematic of the measurement circuit.

5.8 5.9 MAGNETIC FIELD <T>

Fig. 3. A 6453. 2-Sl GaAs quantum Hall step measured to high precision

at 1.2 K, and /= 25.5 fiA.

current less than 10~ 1S A, amplifies the difference-voltage sig- nal. (Note that the potentiometer does not require calibration

in this arrangement.) The current source, potentiometer, and

electronic detector are all battery operated. Thermally induced

voltages and linear drifts in the current source and the potentiometer are cancelled by reversing the current through the device and the reference resistor. A series of reversals in the order + - - + is made for each of two measurements of VH

which bracket in time one measurement of VR in order to obtain a single data point.

Fig. 3 shows a high resolution mapping of the i = 4 Hall step

for one of the GaAs devices cooled to 1.2 K with / = 25.5 juA. Each data point was obtained in one hour using the procedure

described above, with a ±0.011 ppm random, or type A, measurement uncertainty. This Hall step is flat to within ±0.01 ppm over a range in magnetic field that is 2 percent of the

central field value. The i = 4 step of the second GaAs sample was equally as flat, had a nearly identical shape, and occurred

at 6.02 T central field. Both devices clearly have Hall step shapes that make them suitable for use as resistance standards.

III. Quantum Hall Effect Results

One of the room temperature, 6453.2 £2, reference resistors was compared with two different Hall probe sets of each of

the two GaAs devices for both magnetic field directions over a 12-month time period starting in May 1983. The results are shown in Fig. 4, where AR/R = {VH

- VR )j VR . For a measurement time of 1 h the data typically had a ±0.011-ppm

random uncertainty, and was corrected for a measurement sys- tem offset error [3] which was sometimes as large as (0.025 ± 0.013) ppm. This offset error was determined by replacing the

Hall device by a second room temperature 6453.2 ft resistor and then intercomparing the two resistors with the measure- ment system. Also, a correction for the temperature depen160
CAGE etai: QUANTUM HALL EFFECT AS RESISTANCE STANDARD

1 1 1 - r r r it'll

i i

1

0 I 23456789 10 1112

TIME IN MONTHS — STARTING MAY I , I 983

Fig. 4. Relative comparisons as a function of time of a 6453. 2-fi room

temperature reference resistor with two different quantum Hall devices.

\ 0.02 ppm H

0 I 2 3 4 5 6 7 8 9 10 II 12

TIME IN MONTHS — STARTING MAY I . I 983

Fig. 5. Monitoring as a function of time the value of Rh expi ^ssed as a difference in ppm from a nominal value of 6 r53.2 ^nbSdence [4] of RH for each sample was applied to every measure- ment, the largest correction being (0.026 ± 0.002) ppm. No

current dependence [5] was observed for I < 25.5 juA, so no

correction for finite I was required. Our one standard devia- tion uncertainty, the root-sum-square of the above three uncertainties, is ±0.017 ppm. The corrected data were independent of the Hall device, the

Hall probe set, and the magnetic field direction, so they are not distinguished in Fig. 4. From a least squares fit, these data

indicate that the value of the reference resistor is decreasing at the rate of (1.81 ±0.46) parts in 10 10 per day or (0.066 ± 0.017) ppm/year.

IV. Step-Down to the NBS Ohm

The second part of the experiment consisted of calibrating the 6453.2-ft resistors in terms of the set of five one ohm re- sistors which comprise the NBS ohm. To carry out this calibra- tion we constructed two 6453.2 to 100 J2 series-parallel Hamon

resistor networks [6] consisting of eight 800-J2 resistors plus a series-connected 53.2-J2 resistor. The eight 800-J2 resistors in the parallel (100 £2) configuration, as well as the 53.2-12 resistor, were each compared with the series (100 J2) configuration of an existing 100 to 1 J2 Hamon resistor network using a dc

current comparator resistance bridge. The 53.2-12 resistor need only be measured within 1.2 ppm to achieve a 0.01 ppm

accuracy for the 6453.2-J2 Hamon network. The transfer from 1 S2nbs to 6453.2 S2 is estimated to be accurate to within 0.044 ppm (one standard deviation) by this method.

Measurements involving the entire sequence (quantum Hall

resistance compared with the 6453.2-J2 resistors and then stepped-down to S2nbs) have been made since August 1983.

Fig. 5 shows the results of these measurements. The total one

standard deviation root-sum-squared uncertainty of this se- quence is typically ±0.047 ppm, and the mean value of/?// is (1.627 ±0.047) ppm larger than a nominal value of 6453.2

£2n BS . (An uncertainty of 0.047 ppm was assigned to the mean value because the resistance step-down errors are mostly

systematic, or type B.)

V. Conclusions

Although the precision and reproducibility of these measure- ments is 0.047 ppm, the inaccuracy with respect to the SI ohm

is at least 0.5 ppm due to possible drifts of the NBS ohm since

it was determined in SI units in 1974 via a calculable capacitor

experiment [7]. Also, many factors such as the temperature

[4] and current [5] dependencies must be closely examined

and understood before we can confidently use the quantum

Hall effect as an absolute resistance standard. Clearly, however, quantum Hall resistance devices can be used to monitor

or maintain a laboratory unit of resistance to the level of accu- racy required at national standards laboratories. Acknowledgment

The authors wish to thank A. C. Gossard of AT&T Bell Laboratories for growing the GaAs devices, D. C. Tsui of Princeton University, Princeton, NJ, for preparing and screening the devices, and B. N. Taylor for his continued support and

encouragement.

References

[1] K. v. Klitzing, G. Dorda, and M. Pepper, "New method for high- accuracy determination of the fine-structure constant based on

quantized Hall resistance," Phys. Rev. Lett., vol. 45, pp. 494-497,

Aug. 1980.

[2] D. C. Tsui, A. C. Gossard, B. F. Field, M. E. Cage, and R.F.Dziuba,

"Determination of the fine-structure constant using GaAs-Al^- Gai_xAs heterostructures," Phys. Rev. Lett., voL 48, pp. 3-6,

Jan. 1982.

[3] M. E. Cage, R. F. Dziuba, B. F. Field, R. J. Wagner, D. C. Tsui, A. C. Gossard, and N. N. Tadros, "Progress report on investigations of the quantum Hall effect as a possible resistance standard," Com.

Consult. d'Electricite, Com. Int. Poids Mes., Trav. 16e

Sess., Doc. CCE/83-32, Mar. 1983.

[4] M. E. Cage, B. F. Field, R. F. Dziuba, S. M. Girvin, D. C. Tsui, and

A. C. Gossard, "Temperature dependence of the quantum Hall resistance,"/^*. Rev. B., vol. 30, pp. 2286-2288, Aug. 1984.

[5] M. E. Cage, R. F. Dziuba, B. F. Field, E. R. Williams, S. M. Girvin, A. C. Gossard, D. C. Tsui, and R. J. Wagner, "Dissipation and dy- namic nonlinear behavior in the quantum Hall regime," Phys. Rev.

Lett., vol. 51, pp. 1374-1377, Oct. 1983.

[6] B. V. Hamon, "A 1-100 ohm build-up resistor for the calibration of standard resistors," /. Sci. Instrum., vol. 31, pp. 450-453, Dec.

1954.

[7] R. D. Cutkosky, "New NBS measurements of the absolute farad and ohm," IEEE Trans. Instrum. Meas., vol. IM-23, pp. 305-309,

Dec. 1974.

161
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985 A High-Accuracy Automated Resistance Bridge for

Measuring Quantum Hall Devices

BRUCE F. FIELD, member, ieee

Abstract-An automated resistance bridge has been constructed specifically to measure the Hall resistance of semiconductor devices which

exhibit the quantum Hall effect. The bridge is used to perform a oneto-one comparison of the Hall resistance to a reference resistor of similar value. A measurement accuracy of 0.01 ppm or better is expected.

I. Introduction

THE QUANTUM Hall effect is a phenomenon whereby

certain semiconductor devices cooled to near absolute

zero and placed in a high magnetic field (=«6 T or greater) produce a resistance, at small currents, that is quantized and

in theory is related to fundamental constants via

h 25 812.8 Rh = —* : «

e i i

where RH is the Hall resistance, h is Planck's constant, e is the

electron charge, and / is an integer quantum number [1] . To

investigate fully the quantum. Hall effect requires a very large number of high-accuracy resistance measurements. Present quantized Hall resistance measuring systems generally require constant attention by an operator [2] -[5] . A fully automated

resistance bridge, which consists entirely of room temperature

instrumentation, was developed to measure quantum Hall re- sistances in the range 3-13 k£2 with a measuring current adjust- able from 0 to 37.5 /uA. The bridge has been constructed with

a ratio of one-to-one and will be used to compare with high accuracy the resistance developed in the quantized Hall devices

to a room temperature reference resistor of nearly the same

value. Such comparisons will allow one to monitor a unit of

resistance and to investigate the fundamental characteristics of quantum Hall semiconductor devices. Comparison of the (oddvalue) room temperature reference resistor to the as-maintained ohm at the l-O level can be accomplished using a number of methods, including room temperature current comparators,

cryogenic current comparators, or series-parallel resistance networks [6]

II. Bridge Design

A simplified schematic of the bridge is shown in Fig. 1 . Three

electronic detectors (D 1 -D3 ) are used to compare the voltage

drop across the semiconductor device (Hall bar) to a room

Manuscript received August 1984. This paper was supported in part by the U.S. Office of Naval Research. The author is with the Electricity Division, National Bureau of Standards, Gaithersburg, MD 20899.

Fig. 1. A simplified diagram of the measuring bridge. Di~D 3 are elec- tronic detectors with isolated outputs read by digital voltmeters.

temperature reference resistor (R s) constructed to be within a few ppm of the Hall resistance. The isolated outputs of the

detectors are fed to three digital voltmeters which are read simultaneously by a desktop computer. Each detector is powered by a separate battery supply to ensure isolation be- tween the detectors, the ac power line, and ground. The cur- rent source consists of four mercury batteries and a number of

resistors that may be selected to provide a device current be- tween -37.5 and +37.5 juA in steps of 2.5 /iA. Resistors Rl

and R2 are approximately 6.4532 kft and are closely matched

to each other; however their ratio is not exactly known. The

interchange switch exchanges the position of Rl nndR2 in the

bridge so that the effect of the mismatch of R\ and R2 may

be eliminated from the measurement.

At the voltage inputs of the detectors there are no switches which could produce thermal EMF's during the operation of

the bridge. British Post Office (BPO) connectors are used to connect the Hall bar and the reference resistor to the bridge so that their position in the bridge may be exchanged to test for systematic errors. This also provides generality, enabling any

pair of probes on the Hall bar to be measured. The small lon- gitudinal voltage along the device (Vx ) can be measured by

connecting either Z), or D2 (depending on the probe set and

magnetic field direction) directly to the Hall bar and using the detector as a voltmeter. This measurement of Vx is required

to permit extrapolation of the Hall resistance to zero temperature [7]

162
FIELD: MEASURING QUANTUM HALL DEVICES

HI. Guarding

Not shown in Fig. 1 is the circuitry which is used to guard

all the critical bridge components and wiring. Coaxial cable is used for all wiring within the bridge, with the shield of the

cable being driven at the same voltage as the inner conductor. Connectors, switches, resistors, and detectors, etc., are mounted

within shielded boxes which, so far as possible, are driven at the same voltage as the internal circuitry. These shields are driven by a battery source identical to the main current source

so thai although the two sources are completely isolated from

each other, their outputs track one another with temperature and load current. Of particular interest are the computer-operated relays used

to implement the resistor interchange switch, the current re- versing switches (both main and guard currents), and the cur- rent magnitude selection switches. Each relay consists of a read-relay surrounded by a continuous copper shield on which

the actuating coil is wound. The copper shield is extended as part of the bridge guarding circuitry so that leakage currents generated by the actuating coils are shunted away from the

critical cricuitry. By design, thermal EMF's generated by the

relays do not affect the bridge reading. At rated power, however, the relays produce thermal EMF's large enough to over- load the detectors. Consequently, the relays are actuated with

an initial current pulse which is reduced to approximately 25

percent of rated power after 100 ms. This typically reduces

the thermal EMF's to 1 /iV or less.

IV. Detectors

Three guarded, electronic detectors, powered by separate, guarded, battery supplies, are used to monitor the bridge unbalance simultaneously. The three detectors have transformerisolated outputs that feed the floating inputs of three 5-i digit voltmeters. The detectors are operated near null except for D2 which measures the entire voltage difference {R H - Rs)I,

where / is the current through the resistors. The input resis- tance of each of the detectors was measured and found to be 38 Mfi or more, even when off null. This causes negligible loading (<0.0013 ppm) when the Hall resistance RH is within ±10 ppm of the reference resistor Rs

. The calibration of the gain of the detectors is done in situ by inserting in turn, a small adjustable calibrated voltage source in series with each of the detector inputs. The day-today gain uncertainty of the detectors is mostly random and

typically 0.024 percent (one standard deviation estimate or

la). For especially large differences between RH and Rs the

calibrated source is inserted in series with detector D2 at C

during the measurements to reduce the detector voltage to near zero.

V. Bridge Operation

The computer is programmed to adjust the current source to the desired bridge current and to set the interchange switch to the "up" position. The small trimmer resistors at A and B are manually adjusted to minimize detector readings D, and £> 3

. (The guard voltage is also manually adjusted temporarily using detector DY to monitor the difference between the guard voltTABLE I Difference Between Two Resistors in ppm as Measured with the Manual and Automated Systems

New Automated

Date Manual System Bridge

4/19/84 0.471 ±0.011

4/18/84 0.462 ± 0.016 4/18/84 0.450 ± 0.015 6/1/84 0.449 ± 0.008 6/25/84 0.423 ± 0.008 6/25/84 0.454 ± 0.011 6/26/84 0.429 ± 0.010 6/27/84 0.441 ± 0.013 Mean 0.450 ± 0.011* ppm 0.447 ± 0.012* ppm

Difference between the manual

and automated systems 0.003 ± 0.016 ppm

*These uncertainties are the combined la estimates of the random

uncertainties for each measurement.

age and the common-mode detector voltage.) The outputs of

the three detectors are then simultaneously integrated for 30 s. The current polarity is reversed and the detector outputs are again integrated. The interchange switch is set to the "down"

position and two more measurements are made, first with re- versed current and then with the normal current polarity. The

current reversal pattern (+, -, -, +) eliminates errors due to both thermal EMF's and detector offset voltages that either remain constant or drift linearly with time. The pattern is re- peated again with the interchange switch first in the "down"

position and then in the "up" position. This pattern also tends

to eliminate errors due to linear drift in the ratio of resistors Rl and/?2.

The interchange and current polarity switches are implemented using four independently controlled single-pole reed

relays described earlier, and are operated as make-before-break and break-before-make switches, respectively. Current reversal and resistor interchange can thus be performed with almost no

transient voltages appearing at the detectors, and measurements

can begin within a few seconds after switching. All relays are

in the bridge current paths or are in series with large resistors so thermal EMF's in these switches will not introduce an error

in the measured resistance ratio so long as the detector outputs

are measured simultaneously. The three digital voltmeters that read the outputs of the detectors are programmed to respond

to a single trigger command from the computer and then inte- grate the outputs for the same length of time.

VI. Results

Comparisons of two 6453.2-12 resistors were made using the automated resistance bridge and a manual resistance measure- ment system previously used for quantum Hall resistance measurements [5] . Table I summarizes the results. The uncertainties given for all the comparisons are the la estimates of the random uncertainty only for a measurement time of two hours. Despite an apparent drift in the resistors which is inconsistent with previous data, the results are in good agreement.

The automated bridge was also used to compare a quantum

Hall device to a 6453.2-J2 resistor. This resistor had been cali- brated previously in terms of the NBS ohm. Eight measure163
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

merits were made during a one week period with the result RH = 6453.2[1 + (1.1594 ± 0.047) X 10"6

] J2 NBS where a la

uncertainty of 0.044 ppm has been included for the step-down

from 6453.2 fi to 1 fi NBS . The value &RHIRH = (1 .594 ± 0.047) ppm is the good agreement with previous manual system

determinations which gave ARHjR H = (1 .624 ± 0.047) ppm

[8] . Unfortunately, problems that developed with the manual

system prevented a simultaneous comparison of the two sys- tems using a quantum Hall device. The bridge was tested to verify that thermal EMF's in the

relays would not contribute an error to the resistor comparison

by measuring the difference between two resistors using zero

bridge current on several different days. The difference, which

should have been zero, was found to be -0.013 ± 0.008 ppm.

This result is not statistically different from zero at a 90-per- cent confidence level (±0.018 ppm). Another test was per- formed to estimate the sensitivity of the bridge to the setting of the trim resistors at A and B in Fig. 1 . These were offset in random combinations equivalent to a 30-ppm resistor differ- ence and within statistical fluctuations no change was found in the actual measured difference.

VII. Conclusions

The automated bridge has been used successfully to measure

the resistance of a quantum Hall device. Using three detecto s simultaneously has introduced no major error or problems in measuring such a device. Preliminary comparisons using both

resistors and GaAs quantum Hall devices have indicated agree- ment with our manual measurement system to within 0.01 ppm.

Nevertheless, more testing needs to be done in order to gain

full confidence in the new system and to evaluate completely

its systematic uncertainty. Specifically, the short-term random

measurement error is somewhat larger than expected, probably due to a noisy detector, and there is an as yet unexplained,

systematic error in the bridge of 0.018 ppm that can be observed when the positions of the Hall device and reference

resistor are exchanged in the bridge.

We emphasize that the bridge is fully automated and will run completely unattended.

Acknowledgment

The author wishes to thank A. V. Gredone for help in constructing the bridge and S. D. Lince for making the tedious

resistor comparisons with the manual measuring system.

References

[1] K. von Klitzing, G. r>orda, and M. Pepper, "New method for high- accuracy determination of the fine-structure constant based on

quantized HaU resistance," Phys. Rev. Lett., vol. 45, no. 6, pp. 494-497, Aug. i980.

[2] A. Hartland. "Use of a cryogenic current comparator to determine

the quantized Hall resistance in a silicon MOSFET," in Precision Measurement and Fundamental Constants II, B. N. Taylor and W. D. Phillips, Eds., NBS, Spec. Publ. 617, USGPO, Washington,

DC, Aug. 1984, pp. 543-548.

[3] E. Braun, P. Gutmann, G. Hein, F. Melchert, P. Warnecke, S. Q. 7 je, and K. v. Klitzing, "Cryogenic method for the determination

jf the fine-structure constant by the quantized Hall resistance," in Precision Measurement and Fundamental Constants II, B. N. Taylor and W. D. Phillips, Eds., NBS, Spec. Publ. 617, USGPO, Washington, DC, Aug. 1984, pp. 535-538.

14] G. W. Small, "Comparison of quantized Hall resistance with a 1 n

standard," IEEE Trans. Instrum. Meas., vol. IM-32, pp. 446-447,

Sept. 1983.

[5] T. Endo, Y. Murayama, M. Koyanagi, J. Kinoshita, K. Inagaki, C. Yamanouchi, and K. Yoshihiro, "Measurement system of quantized

Hall effect by utilizing a Josephson potentiometer," pp. 323-327,

this issue. [6 ] D. C. Tsui, A. C. Gossard, B. F. Field, M. E. Cage, and R. F. Dziuba, "Determination of the fine-structure constant using GaAs-Alx Gaj_x As Heterostructures," Phys. Rev. Lett., vol. 18, no. 1, pp.

3-6, Jan. 1982.

[7] M. E. Cage, B. F. Field, R. F. Dziuba, S. M. Girvin, A. C. Gossard, and D. C. Tsui, "Temperature dependence of the quantum Hall re- sistance," Phys. Rev. B, vol. 30, no. 4, Aug. 1984.

[8] M. E. Cage, R. F. Dziuba, and B. F. Field, "A test of the quantum

Hall effect as a resistance standard," pp. 301-303, this issue.

164
1982 Annual Conference

A POSSIBLE QUANTUM HALL EFFECT RESISTANCE STANDARD

M. E. CAGE, R. F. DZIUBA, and B. F. National Bureau of Standards

Washington, DC

FIELD

ABSTRACT

The discovery of the quanta Hall effect by K. V. Klitzing, using semiconductor devices that are cryogenically cooled in large applied magnetic

fields has opened up the exciting possibility that this effect could stimulate the discipline of electrical metrology to an extent analagous to that of the Josephson effect. This paper will describe

the quantum Hall effect in an attempt to achieve a new resistance standard accurate to a few parts in 1Q&, in which the resistance is defined in terms of fundamental constants of nature.

INTRODUCTION

This paper describes a new phenomenon that

promises to have the same kind of impact on electrical metrology that the Josephson

effect has had. It is called the quantum

Hall effect, and it uses specially made

semiconductor devices that are cryogenically cooled to temperatures near absolute

zero. They are placed in a large magnetic

field produced by a superconducting magnet

in order to achieve a two-dimensional elec- tron gas in the semiconductor devices. The

electrons in this two-dimensional gas can

then be made to completely occupy all of

the quantum states characterized by a known

integer number i. Whenever this condition

is achieved the resistance v of the device, defined by the ratio"of the Hall

voltage VH across the device to the current

I through it, is found to have discrete,

reproducible values that satisfy the relationship .

25,812.80

e

2 i (1)

Rj, is the quantized Hall resistance, and

its value depends upon the quantum integer

i, Planck's constant h, and the electron

charge e. Thus we have a resistance that

can be determined solely in terms of fundamental constants of nature. This astonishing result was first verified

by Klaus vonKlitzing* in West Germany about

two years ago. He demonstrated that the

quantized Hall resistance satisfies Eq . (1) to at least the 10 ppm ( par ts-per-mi 1 1 ion

)

level of accuracy. His work has generated a large amount of excitement in the scientific community, and national metrology

laboratories all over the world are starting to investigate this phenomenon.

MOTIVATION

Why are resistance metrologists so interested in this effect when they already have

high quality, stable, wire-wound resistance

standards? The reasons are two-fold.

First, as is well known, wire-wound resistors do not travel well: their values can change due to mechanical and thermal

stresses, as well as barometric pressure

variations, during transport. The limited

data available from international comparisons indicate possible resistance transport

shifts in the range 0.05-0.2 ppm for 1 ohm

resistors, and significantly larger shifts

for higher value resistors. Second, these

resistors are artifacts and cannot a priori

be expressed in "absolute" (i.e. Systeme

International or SI) units. In the United

States, for example, the as-maintained unit

of resistance is defined to be the average

of five nominal value 1 ohm resistors that are kept at a 25°C mineral oil bath at the

NBS . This does not imply, however, that the

U.S. unit of resistance is not known in terms of the SI unit. It has been so determined via an extremely difficult measurement

based on the NBS calculable capacitor.

2

This measurement was last performed in 1974

and because of its difficulty is only now

being repeated. The one standard deviation

uncertainty of 0.03 ppm achieved in 1974 is

3 times better than any other attempt. This

means ' that if the quantized Hall resistance

is to be used as an SI resistance standard,

von Klitzing 's original work must be pushed

about three more orders of magnitude. We believe that we are now within about a factor of ten of achieving this goal. But do

not be alarmed: wire-wound resistors would

continue to be used as working standards,

with quantum Hall effect resistance standards being maintained in national metrology

laboratories and perhaps eventually also in

a few primary laboratories. They would be

used in much the same way as the Josephson

effect is used to maintain the unit of

voltage: the national wire-wound resistance

standards would periodically be calibrated

against the quantized Hall resistance in order to know their time dependences and to assure that they are consistent with the SI units

.

QUANTUM HALL EFFECT

With the above as an introduction and the

motivation for doing this experiment, let us now investigate the quantum Hall effect in more detail. We first need a suitable semiconductor device. There are two device

types which have been successfully used:

laboratory versions of silicon MOSFETs

( metal-oxide- semi conductor field effect

165
1982 Annual Conference

transistors) and GaAs-Al xGa^- x As heterojunctions (which are hard to fabricate; to date, no more than a dozen good GaAs

hetero junctions have ever been made). We will only briefly describe how our Si

MOSFETs are made at the Naval Research Laboratory in a collaboration with F. J. Wagner. We start (see Fig. 1) with a single crystal substrate of p-type silicon

and grow a thin SiC>2 layer on the top surface for electrical insulation. A metal

film of gold or aluminum is then evaporated

over the SiC>2 layer to form a gate and pads

for the source and the drain (which electrically contact the substrate via diffused

regions of heavily doped n-type material).

The devices (see Fig. 2) are 1.3 mm long

and 0.2 mm wide and can be seen with the

naked eye. In addition to the source and

drain pads we have potential probes placed

along the device which sample voltages in the channel under the gate.

We ground the source (see Fig. 3) and apply

a positive dc voltage to the gate, thereby

obtaining a parallel-plate capacitor, with

the metal gate being one electrode and the

semiconductor substrate the other. The

Si02 layer is typically 0.1pm thick, so if

the gate voltage is 30 volts this means

that the electric field across the insulator is 3 x 10^ volts/cm! Therefore, no pinholes can be tolerated in the Si02

layer. By increasing the gate voltage we induce more electrons near the top surface

of the substrate. But it is not like a normal capacitor because these electrons

are bound in a potential well and must therefore obey quantum mechanical laws.

This potential well quantizes the electrons

in the vertical direction and con fines them

into a two-dimensional electron gas. The

gas is so ner the Si02 interface that this

surface must literally be atomically

smooth, otherwise the electron gas will

scatter from the rough surface and destroy

the quantum Hall effect. Also, there are always impurity ions present and the electrons will scatter from them; we therefore

require state-of-the-art devices to minimize these problems.

By applying an electric field in the vertical direction to a sample cooled to near absolute zero we have quantized the electrons in the vertical direction, forming a two-dimensional electron gas. Let us next apply a magnetic field that is also in the

vertical direction. This must be a very

large field, so we use super-conducting

magnets to produce fields up to 15 T, which

is 150,000 gauss (or about 3 x 10^ times

larger than the Earth's magnetic field).

The electrons in the two-dimensional gas

form cyclotron orbits about the vertical

magnetic field lines, quantizing the electrons in the horizontal plane. The result

is a completely quantized, two-dimensional

electron gas. By increasing the gate voltage we induce more electrons into allowed

cyclotron orbits, and it is possible to completely fill, for example, all of the

i = 4 orbits, but none of the i = 5 orbits

because the device is so cold, and the current is so low, and the scattering is so small that there is insufficient energy to occupy any i = 5 states.

Now assume that we have just filled the

i = 4 states and then pass an electron current through the sample. There is the

resulting flow pattern (see Fig. 4): the

electrons emerge from one corner of the

device and exit at the opposite, the reason being that magnetic forces cause the electrons to pile up in the upper left hand

corner and along the top edge. The magnetic forces are exactly balanced by the

electrical forces between the electrons

within the sample, but there is an escape

route at the top right hand corner. The

electrons are acclerated at the ends of the

device and drift with constant velocity in

the interior. If the current is 5 u A and

the i = 4 quantum states are just filled,

then the source- drain voltage is about 32 mV . Let us next map the potential distribution around the periphery of the sample (as

shown in Fig. 4) with the source at zero

volts since it is grounded. We can measure voltages within the sample via the potential

probes. For example, the voltage drop along

the channel, V x can be obtained with probe

sets 1,2 or 3,4 and in both cases V x = 0. The reason is simple: if the i = 4 states

are filled and if there is insufficient

energy to scatter into any i = 5 states,

then there is no scattering. If there is no scattering there is no voltage drop. Thus

the voltage drop along the channel becomes

very small when all the quantum states are filled. There is, however, a voltage drop across the channel, V v = Vjj , and this Hall

voltage can be measured with probe sets 1,3 or 2,4. In both cases VH

is about 3 2 mV

K. v. Klitzing measured more carefully-*-

and found in sample after that VH was 32.266

mV. Divide this by 5 A and you get = 6,453.2 Q, which is precisely the value predicted in Eq. (1) for i = 4!

MEASUREMENT SYSTEM

Can this amazing result be pushed the three

more orders of magnitude necessary for use

as an SI resistance standard? Here is the

measurement system (see Fig. 5) that we are using in our initial attempts to answer

this question. The system is batteryoperated an<l consists of a constant current

course that supplies current to the Hall

device which is cooled to 4.2 K by immersing the device in liquid helium. We can further cool the sample of 1.5 K by pumping

on the liquid helium bath. (The Hall

sample is, of course, located in the central field of a superconducting magnet.)

The current also passes through a room tem- perature reference resistor constructed to have the same nominal resistance value as the quantum Hall resistance of interest,

e.g. 6,453.20 Q when i = 4. This assures us that VH Z VR, and that both voltages can be

nearly bucked-out by the potentiometer. The

electronic detector, D, reads the voltage

166
1982 Annual Conference

differences and its isolated output is

either directed into an X-Y recorder or is

digitized for computer analysis. If we can assure ourselves that the same current

passes through both the sample and the ref- erence resistor, then the current can be

eliminated in the equations for VH and Vr

and the quantized Hall resistance RH can be

obtained from the ratio ^h/Vr and the calibrated value of the reference resistor RR.

(RH will be given in Si units if RR has

been determined in SI units via the calculable capacitor experiment. ) 2 The cur- rent is typically 10 u A., so for a part in

1Q8 measurement this requires that any cur- rents shunting the sample or resistor, or leaking to ground, must be less than

10~13a. Another way of stating this condition is that the leakage resistance between

the leads, and to ground, must be greater

than lO 1-^ fi --except for the gate lead,

which must be greater than 10*4 fi since it

is at a higher voltage. (A fingerprint

across terminals can be a short-circuit in

this experiment!)

RESULTS

Now for some data. Here are low sensitivity

X-Y recordings of Vh and v x , in mV, versus

gate voltage for a silicon MOSFET at 13 tesla obtained at the Naval Research Laboratory (see Fig. 6). The voltage drop

along the channel, V x becomes very small

when a quantum level is filled, whereas

plateaus or steps simultaneously occur in

the Hall voltage because the electrons have

insufficient energy to occupy cyclotron

orbits with larger quantumnumber s . It is easy to identify the quantum states: for

example, a step occurs at about 32 mVr divide that by 5 g A and we have the

6,453.20 fi step. We have just filled all

of the possible i = 4 cyclotron orbits.

We can investigate this step in more detail

by bucking-out most of the Hall voltage

with the potentiometer and reading the

voltage differences with the electronic

detector. Here is the Hall step on a scale

expanded by 4 orders of magnitude (see

Fig. 7). The arrow indicates the minimum

Vx

. There are large thermally-induced

voltages in the leads between the sample at

1.5 K and room temperature; they are eliminated by obtaining a Hall step of the opposite polarity (by reversing the current)

and by taking half the difference of the two polarities (since the thermal EMFs do

not reverse on current reversal). The

resulting Hall step is flat to within at least 1 ppm over a gate voltage range of

0.2 V. It takes about one minute to obtain

a Hall step plot at this sensitivity, so we

can quickly obtain a lot of information

about its overall shape. For precision

results, we fix the gate voltage and

digitize the data for 1-2 hours.

collaboration with D. C. Tsui and A. C. Gossard. Here are high sensitivity plots

for three Hall steps (see Fig. 8), two for

i = 2 (12,906.40 fi steps) and one for i = 4 (6,453.20 Q ). (In Ga As he tero j unct ions we vary the magnetic field rather than the gate

voltage to change the quantum state filLing

factor.) We believe that the different

shapes on the sides of the steps are due to inhomogenities in the sample.-^ Our result^

for these three samples, expressed in 31 units, is: RH = 6 , 45 3 . 2004 ( 1 1 ) fj for i = 4, with a one standard deviation uncertainty of

+ 0.17 ppm. This result is three to eight

times more accurate than the other three

quantized Hall resistance results obtained

by workers in West Germany ^ and Japan. 6,7

We have also made measurements on silicon

MOSFETs, plus direct silicon MOSFET —Ga As hetero junction in te rcompa r isons , at the

Naval Research Laboratory (NRL), and find

the values of RH to be sample-independent

to at least the 0.1 ppm level of accuracy.

However, much to our surprise, there is a problem because the values of Rh obtained

at NRL were found to be about 0.8 ppm

larger than those we obtained at Bell Laboratories. We currently attribute this

discrepancy to rectification of electrical

noise in the highly nonlinear samples.

This assumption is now being investigated

in a much cleaner electrical environment at NBS using an 8 tesla magnet.

SUMMARY

If the results confirm our Bell Laboratories measurements, we will have advanced

the experiment two of the three orders of

magnitude necessary to achieve an SI resistance standard based upon fundamental con- stants of nature. What about the final

order of magnitude? It will not be easy, but we believe that it is experimentally

possible. A new measurement system is under construction for our second generation experiments, and a 15 tesla magnet

should be delivered vithin a year. There

are certain to be unexpected problems and

surprises along the way but we remain optimistic .

ACKNOWLEDGEMENTS

The authors wish to acknowledge the

encouragement of B. N. Taylor and the

important contributions of our collaborators: S. M. Girvin and N. N. Tadros of

NBS, A. C. Gossard of Bell Laboratories,

D. C. Tsui of Princeton University, and

R. J. Wagner of the Naval Research

Laboratory. The work at NBS is supported in

part by the Office of Naval Research and by

the Calibration Coordination Group of the

Department of Defense.

We have also obtained data on GaAs heterojunctions at Bell Laboratories in a 167
1982 Annual Conference

REFERENCES

K. v. Klitzing, G. Dorda, and M. Pepper, "New Method for High Accuracy

Determination of the Fine-Structure

Constant Based on Quantized Hall Re- sistance," Phys. Rev. Lett. 4_5, 494

(1980)

R. D. Cutkosky, "New NBS Measurements

of the Absolute Farad and Ohm, " IEEE Trans. Instrum. Meaa. IM-23 , 305

(1974)

R. J. Wagner, C. F. Lavine, M E. Cage, R. F. Dziuba, and B. F. Field,

"Measurements of the Quantized Hall

Steps in Si at the ppm Level," Surface

Sci. 113, 10 (1982).

D. C. Tsui, A. C. Gossard, B. F. Field, M. E. Cage, and R. F. Dziuba,

"Determination of the Fine-Structure

Constant Using GaAs-Al xGa l-xAs Heterostructures," Phys. Rev. Lett. 48, 3 (1982)

NRL n -CHANNEL SILICON MOSFET

(TOP VIEW)

POTENTIAL PROBES

SOURCE \ /

POTENTIAL PROBES DRAIN

.3 mm

Figure 2

E. Braun, E. Staben, and K. V. Klitzing, "Experimental determination

of h/e^ by the Quantized Hall Resistance in MOSFETs , " PTB Mitteilugen 90 , 350 (1980).

SILICON MOSFET (Side View)

C. Yamanouchi, K. Yoshihiro, J. Kinoshita, K. Inagaki, J. Moriyama,

S. Baba, S.Kawaji, K. Murakami, T. Igarashi, T. Endo, M. Koyanagi, and

N.Nakamura, "Hall Effect in Silicon

MOS Inversion Layers - For h/e^ Determination," in Precision Measurement

and Fundamental Constants 1

, ed . by

B. N. Taylor and W. D. Phillips (Nat. Bur. Stand. U.S., Spec. Publ. 617, to appear )

K. Yoshihiro, J. Kinoshita, K. Inagaki, C. Yamanouchi, J. Moriyama,

and S. Kawaji, "Quantized Hall Resistivity in Si-MOSFETs Measured at Liq.

^He Temperatures," J. Phys. Soc . Japan

51, L5 (1982).

+ + + + + + + v/A ty///////// ^w//////////

\ P7%\ D t

p- Si

Figure 3

T

SILICON MOSFET (Side View)

SOURCE

GATE

SUBSTRATE

Figure 1

DRAIN

EQUAL- POTENTIAL LINES AND ELECTRON PATHS

R H

= 6,453.20X1

I so

= + 5^.A

168
1982 Annual Conference

S 10 15 20 GATE VOLTAGE (Volts)

Figure 6

T I I i i i r

J 1 1 l l I L VG (200 mV/div.)

Figure 7

169
The Quantum Hall Effect. I

The recent surprising observation of a quantization of the Hall resistance in units of

hie2 in quasi-two-dimensional conductors presents the possibility of obtaining an improved

value of the fine structure constant and development of a quantum standard of resistance

using a solid state device.

INTRODUCTION

The recently discovered quantum Hall effect has attracted a great deal

of interest due to the possibility of obtaining an improved value of the

fine structure constant 1 and development of a quantum standard of re- sistance. The observation of a quantization of the Hall resistance in

units of hie2 in quasi-two-dimensional conductors has necessitated a major rethinking of our picture of transport in these systems. It is the

purpose of this Comment to provide an introduction to this phenomenon

and an overview of the important questions in the field which are currently being discussed.

We begin with a brief introduction to the subject of electron inversion

layers and the discovery of the quantized Hall resistance. This will be

followed by a discussion of the simple theory for the case of no disorder.

Next we will consider the theoretical and practical implications of an improved value for the fine structure constant and achievement of a quantum mechanical standard of resistance. We will conclude with a

list of questions of specific current interest which will be discussed in

more detail in a later Comment.

Comments Solid State Phys. © 1983 Gordon and Breach

1983, Vol. 11, No. 1, pp. 1-16 Science Publishers, Inc. 0308-1206/83/1 101-0001/$ 1 8. 50/0 Printed in the United States of America

170
INVERSION LAYERS

One of the features of physical systems which is of great importance

is the dimensionality. There has been particular interest in recent years

in seeking out and studying quasi-one- and -two-dimensional systems.

One of the most versatile and successful examples of such systems is the quasi-two-dimensional electron gas generated in the form of so- called inversion layers at semiconductor interfaces. For a recent comprehensive review see Ando et al.

2 The most common technique involves the use of a device known as a silicon MOSFET (metal-oxidesemiconductor field effect transistor). This is a laboratory version of a commercial device whose structure is shown schematically in Figure 1

.

The typical length scale of these devices is 1 mm. The inversion layer

is formed at the Si-oxide interface by application of a positive (in an

Az-type device) voltage (10-50 V) to the gate. Because the oxide layer

is thin, an electric field of order 106 V cm-1is generated which bends

the Si conduction band downward and attracts carriers to the interface.

This is illustrated in the energy level diagram shown in Figure 2. A

very nice feature of these systems is that the carrier density is continuously adjustable by means of the gate voltage.

While it is true that the inversion layer charge forms a thin sheet, it

is not at all obvious that this system should behave as if it were strictly

two-dimensional. This question was first addressed by Schrieffer, who

pointed out that the electric field from the gate confines the electrons

to such a narrow region (25-50 A) that quantum mechanical effects are

essential to the description (see discussion in Ref. 2). The motion in

the direction normal to the interface is quantized into discrete energy

levels called electric sub-bands whose splitting is quite large (~ 20

meV). This energy scale is much larger than any other in the system

and so motion in the normal direction is "frozen out" by quantum

mechanics. This means that for applied frequencies co such that h& <

20 meV the transport properties are those of a mathematically twodimensional system. This idea was confirmed in a Shubnikov-de Haas

measurement by Fowler et al.

3 which showed that the density of states

is indeed two-dimensional.

Another inversion layer system which has been developed is the

GaAs-Al^Gai^As heterostructure4 which is fabricated by molecular

beam epitaxy. Unlike MOSFETs these systems have a fixed carrier

171
172
FIGURE 2 Electron energy level diagram showing the effects of a positive gate voltage.

When the conduction band is bent down below the Fermi level £F , the inversion layer

is formed at the interface.

density and so the magnetic field must be adjusted to achieve the quantum Hall condition discussed below. Although difficult to construct these

devices offer several advantages. The requisite magnetic fields are relatively low (~ 8 T versus ~ 13 T or more for Si). In addition the

electron effective mass is quite low (0.068 me versus 0.2 me for Si) so

that the characteristic magnetic energy is large, allowing one to work

at higher temperatures (~ 4.2 K). Another advantage is the very high

mobility that can be obtained since, unlike MOSFETs, the donors provide the inversion layer charge and can be physically separated from

the active region.

173
THE QUANTUM HALL EFFECT

In 1980 von Klitzing et al.

1 reported that under certain conditions the

Hall resistance of a MOSFET inversion layer is quantized in units of

hie2 where h is Planck's constant and — e is the electronic charge.

Before delving into this remarkable observation let us briefly review

the classical Hall effect. In an ordinary conductor carrying a current

density J the associated electric field E is given by E = pj and points

in the direction of the current flow. Here p is the resistivity. If a magnetic

field Bz is applied normal to the plane, the resistivity becomes a tensor

quantity:

Et = pijJj, (1)

so that a component of the electric field appears perpendicular to the

current and the magnetic field. This is called the Hall field and the offdiagonal part of the resistivity p^ is called the Hall resistivity.

The resistivity tensor is obtained experimentally by establishing a constant current density J in the x (say) direction and measuring the

resulting electric field. Ex determines p^ and Ey determines pyx . In

principle one could equally well establish a fixed electric field and

measure the resulting current. This would constitute a determination of

the conductivity tensor cr defined by

Ji = dij Ej. (2)

The conductivity is, as usual, the inverse of the resistivity but because

these are tensor quantities it is not true that cr^ = l/p^. Indeed, in the

quantum Hall case where the current is almost perfectly perpendicular

to the electric field one has p^ = cr^ = 0. Thus the resistivity and the

conductivity both vanish! This paradoxical result is easily understood,

however. If an electric field is applied no current flows in the direction

of the field. Hence the (diagonal part of the) conductivity vanishes. If

on the other hand a current is applied, no field appears in the direction

of the current. Hence the (diagonal part of the) resistivity also vanishes.

It turns out in this case that the off-diagonal terms are simply related

by P^ = - 1/ov

The Hall resistivity may be calculated in several different ways, both

quantum mechanically and classically, in terms of the Lorentz force

174
acting on the carriers drifting in the magnetic field. The standard result

is -B

Pyx = , (3) ne

where n is the number of carriers per unit volume (or, in the case of

two dimensions, per unit area). An important aspect of this expression

has to do with the dimensionality. Experimentally, the easiest quantity

to measure is the resistance, which is the measured Hall voltage divided

by the applied current. The quantity given in Eq. (3), however, is the

Hall resistivity. For a sample of length L in each dimension the connection between the Hall resistance RH and the Hall resistivity pyx is RH = pyj?-*, (4)

where d is the dimensionality. It is fortunate that in two dimensions the

length factor drops out so that resistance and resistivity are the same

quantity. If this were not the case accurate measurements would entail

length scale determinations which would be essentially impossible to

perform to high enough accuracy.

Since the inversion layer charge density n is relatively low, one sees

a huge Hall voltage (~ 6 mV/(xA). Because the carrier density in a MOSFET is proportional to the gate voltage VG one expects [from Eqs.

(3) and (4)] a smooth hyperbolic dependence of RH on VG . Under

conditions of high magnetic field (~ 13 T) and low temperatures (~ 2

K) startling deviations from this behavior are seen as shown5 in Figure

3. One sees a series of plateaus now referred to as Hall "steps" where

the Hall voltage VH appears to be essentially constant independent of

the gate voltage. The value of the Hall resistance at these plateaus is to a high degree of accuracy (the errors are believed to be less than 1 part in 106

):

Rh = \, (5) e i

where i is a small integer.

This unexpected result is (seemingly) not difficult to understand for

the ideal case of no disorder, no Coulomb interactions and zero tem175
5 10 15 20

GATE VOLTAGE (Volts)

FIGURE 3 Hall voltage (VH) and longitudinal voltage drop (V,) in a silicon MOSFET

at 1 .5 K and 13 T with a constant 5 ^.A source-drain current. Note that the Shubnikov-de

Haas oscillations in Vx are correlated with the plateaus in VH . The 6453.20 Cl plateau

corresponds to h/4e2

. The quantum number i appears in Eq. (5).

perature. If we assume an effective mass Hamiltonian then we have in

the presence of the magnetic field

H - -i- (p 4- eA)2

, (6) 2m*

where it is convenient to take the vector potential A to be in the Landau

gauge:

A = x By. (7)

Quite often in atomic and condensed matter physics we make the approximation of keeping in the Hamiltonian only the term linear in B.

176
Here, however, the magnetic field is so large that the behavior of the

system is dominated by the field and it is essential to retain the quadratic

term. As shown in Figure 4a the energy levels are quantized into discrete

uniformly spaced harmonic oscillatorlike levels En = n hwc

. The magnetic energy is fta>c where coc is the classical cyclotron frequency.

One can think about the discrete quantization of the energy levels in

the following semiclassical picture. As the strength of the magnetic field

is increased the radius of the classical Larmor orbit decreases (for a given particle speed). At high enough fields quantum mechanics becomes important because the circumference of the orbit reaches the

scale of the de Broglie wavelength. The requirement that the orbit

contain an integer number of wavelengths results in the discrete quantization of the energy levels.

CU,c

(a)

(b)

FIGURE 4. (a) Density of states of the ideal two-dimensional electron gas in high magnetic

field and zero field (dashed), (b) Broadened density of states in the presence of disorder. Shading indicates regions of localized states.

177
For a Si(100) inversion layer hmc = 5.8 meV at 10 T and for a GaAs

heteroj unction ho)c =17 meV at 10 T. These energies are much larger

than characteristic thermal energies 0.1 meV at T :£ 1 K). This is essential to the effect because if the chemical potential lies in the gap

between Landau levels then an integer number of levels will be fully

occupied and all higher ones will be essentially empty. The essence of

the phenomenon is that the density of states per unit area per Landau

level is then proportional to the magnetic field:

eB

n = —. (8)

h

Putting this into Eqs. (3) and (4) we have the desired result for the

quantum Hall resistance:

Rh = hie2

. (9)

If i Landau levels are filled this expression is divided by i, yielding the

observed result of Eq. (5).

The sine qua non of this simple result is the areal quantization which

produces a density of states per Landau level that is proportional to the

applied field and independent of the effective mass. The only quantity

which depends on m* is the energy spacing h(x)c

. The first worry that

comes to mind is the validity of the effective mass approximation since

the bands in semiconductors are never perfectly parabolic and, in addition, the magnetic energy is not small compared to the bandwidth.

Onsager's famous argument6 concerning the de Haas-van Alphen effect

for Bloch electrons shows one how to calculate the magnetic energy

levels for an arbitrary band. However, this argument is only valid for

large magnetic quantum number i and cannot be expected to be valid

in the quantum limit where i is small. This is not a problem, however,

since the actual Landau level spacing is irrelevant so long as it is much

greater than kBT. The counting argument that gives the density of states per Landau level proceeds in wavevector space and should not (in the

absence of random impurities) be dependent on the dispersion relation.

Nevertheless, no rigorous proofs along these lines have been developed.

In any case the areal quantization in the ideal case is something of a

red herring since, as we shall see, it does not apply to the case of strong

disorder where some of the states become localized (and is nevertheless correctly quantized).

178
We have already mentioned the lack of length scale dependence of

the resistance in two dimensions. A profound implication of this is that

in the classical case the rneasured Hall resistance is completely geometry

independent. One can have a variable width or even punch holes in the

sample without affecting the Hall resistance. 7 Tsui and Allen8 have

suggested that this classical result is a good point of departure for

forming a picture of the quantum Hall effect. Iordansky9 and Kazarinov

and Luryi 10 have used a similar picture in semiclassical considerations

of percolation effects in two-dimensional systems with a random potential which is slowly varying on the scale of the magnetic length, / = (h/eB) 1/2

. The essential feature in all these cases that makes the Hall

resistance remain constant is that in two dimensions the current not

being carried in the insulating regions is compensated by extra current

flowing in the conducting regions. The extent to which this applies to

the quantum mechanical case with nonsmooth potentials will be discussed briefly below and in more detail in Part II of this series.

SCATTERING AND DISORDER

The most obvious defect of the simple theory presented above is the

assumption of no disorder. It is true that in the weak field Boltzmann

theory there are no scattering corrections to pyx [note that there are corrections of order (o)ct)

2 to <jyx] but there is no reason to expect this

result to hold in the present case. We know in fact that the disorder

must be quite large because of the anomalously large width of the Hall

steps. If there were no disorder then there would be no states in the

gap between Landau levels and the chemical potential could not be

sustained in the gap for a finite range of magnetic field or gate voltage.

The current belief is that there is sufficient disorder (or a combination

of disorder and Coulomb interactions) to produce a continuous density

of states with no gap. States close to the original Landau level are extended and those further away are expected to be non-current-carrying

(zero mobility) localized states (see Figure 4b). Thus, there is a mobility

gap even though there is no gap in the density of states. This allows

the system to keep the chemical potential in the mobility gap over a

finite range of electron density or magnetic field. Of course the number

of extended current carrying states no longer satisfies Eq. (8) and there

is no reason to expect the Hall resistance quantization. The paradox is

179
that the quantum Hall effect cannot be observed without disorder, and

yet there is no obvious reason to expect it to work with disorder. Somehow extra current must be carried by the extended states to compensate

for the lack of current in the localized states. This point will be discussed

further in Part II. We will touch only briefly here on one point about the disorder which

will show why this system is unique and give a hint as to why disorder

does not destroy the quantization. In an ordinary inversion layer system

unavoidable disorder produces a finite relaxation time t ~ 10-11

s. In

the presence of low temperature and a high magnetic field, however,

the situation is drastically altered if the chemical potential lies in the

mobility gap. All the current-carrying states below the Fermi level are

filled and, just as in a superconductor, elastic scattering of the carriers

cannot occur because of the finite gap. Referring to Figure 3 we see

that there are indeed large (Shubnikov-de Haas) oscillations in the

diagonal component of the resistance (Rx = VJI) and that this resistance

is very small in the region corresponding to each Hall step.

The resistance becomes so low in fact that it is difficult to measure. A lower limit for the high B field relaxation time t* is 108 t ~ 1 ms.

11 The primary limitations in the measurement at present are finite temperature and finite (nonlinear) electric field effects. Measurements of

these effects are of great interest since they provide information about

the nature of the transport. The scattering rate is so small that the

equivalent three-dimensional resistivity is 5 X 10~13 fl-cm 11 which

is lower than any other known value except those of superconductors.

The resistivity is so low that it might be easier to find the relaxation

time directly by measuring the persistence time of current in a closed

loop. This can be done for superconductors but would be very difficult

in these small devices because the stored energy is very small and in

any case is largely capacitive (associated with the applied Hall voltage)

rather than inductive. 11 Another important measure of the scattering rate is p^/p^ = tan 8H , the tangent of the Hall angle. The Hall angle is defined to be the angle

between the electric field and the current, and from the upper limit on

p^c one finds tan 8H ^ 10 10, indicating that the dissipation is indeed

very low and the current flow very nearly perpendicular to the applied

field.

In summary, the peculiar properties of two-dimensional Landau level

systems cause the transport to be dissipationless even in the presence

180
of strong disorder. Thus in principle p« = 0 as in the ideal zero- disorder limit. This is not a proof that p^ should be unaffected but is a strong hint that there is a fundamental connection between this result

and the quantization of the Hall resistance. This connection will be

explored in greater detail in Part II.

THE FINE STRUCTURE CONSTANT AND A POSSIBLE

RESISTANCE STANDARD

The fine structure constant a is the dimensionless measure of the coupling between matter and the electromagnetic field. It is central to the

theory of quantum electrodynamics (QED). The fine structure constant can be related to the quantum Hall resistance by

\iQc e

2 poc i a = —-h = —YH

' (10)

where \xq, the permeability of the vacuum, is by definition exactly 4tt

x 10~7 Him, and c is the speed of light. Because c is very accurately

known (and in fact because of a forthcoming redefinition of the meter

is soon to be an exactly defined number) one can use the quantum Hall

effect (QHE) measurement of RH = hle2

i to determine a. Improvements

in our knowledge of a are important as a check on the internal consistency of QED and, if sufficient accuracy is obtained, can also put

constraints on modem theories of the weak and strong interactions.

The most accurate experimental determination of a to date is the

QED theory-dependent result from measurements of the anomalous

magnetic moment of the electron, 12 ae , with an uncertainty of 0.07 ppm

(parts-per-million). One would like to have a value of a that is independent of QED, and this has been obtained by a combined Josephson

effect 13 (2e/h) - proton gyromagnetic ratio 14 (7P) experiment with an

uncertainty of 0.11 ppm. The quantum Hall effect experiment also

provides a QED-independent value of a, and the accuracies of reported

results are approaching the best values: 1.3 ppm, 15 0.88 ppm, 16 0.50

ppm,

17 and 0.17 ppm.

18 Measurements at these levels of accuracy require access to a carefully

calibrated set of electrical units. The QHE .result for a with the 0.17

ppm uncertainly 18 and the combined lelh - 7P value 13,14 for a were

181
both obtained in terms of the electrical units maintained by the U. S.

National Bureau of Standards. It is possible to combine these two re- sults for a in a way which eliminates much of the uncertainty in the calibration of the ohm and thereby obtain a more precise value 18

: a

-1 =

137.035965(12) (0.09 ppm).

The QHE values of a are in agreement with the ae and 2elh — 7P values, within their uncertainties, and seem to be sample independent

for both Si MOSFETs 1719 and GaAs-AUGa^As heterojunctions. 1819

It seems reasonable to expect that future QHE experiments will provide

even more accurate values of a. We now turn to the question of a possible resistance standard. As

we have seen the quantity hie2 has the units of electrical resistance and

has a value which is in a convenient range for measurement purposes:

hie2 ~ 25 812.80 fl. If the quantized Hall resistance RH is shown to be sample independent, and if it satisfies the simple relationship given

by Eq. (5), then one could use the QHE to achieve an electrical resistance

standard in which the resistance is determined in terms of fundamental

constants. One advantage of this would be facilitation of international

comparison of resistance standards. The present technique involves

transport of wire-wound resistors whose resistance values can change

due to mechanical and thermal stresses during transport. The limited

data available from international comparisons indicate possible resistance transportation shifts in the range 0.05-0.2 ppm for 1-ohm resistors, and significantly larger possible shifts for higher value resistors.

A fundamental problem from the metrology point of view is that the

quantities in Eq. (5) are in "absolute" (i.e., Systeme International or

SI) units. The experimentally determined resistance is not RH but

which is measured in terms of an as-maintained unit of resistance, H*,

and must therefore be converted to the SI unit, O, via the equation

In the United States, for example, the as-maintained unit of resistance

fl* = Hnbs is defined to be the average of five nominal value 1-ohm

resistors that are stored at the National Bureau of Standards in a 25 °C

mineral oil bath regulated to within several m°C. Other 1-ohm resistors can be calibrated relative to these as-maintained resistance standards to a one standard deviation accuracy of about 0.01 ppm using a current

(11)

182
comparator bridge. 20,21 The ratio ft^s/^ has been determined to within

0.03 ppm via a calculable capacitor. 22,23 This extremely difficult measurement is about three times more accurate than any other ft*/ft determination, 24 and it means (at least in the U.S.) that if RH is to be used

as an absolute resistance standard, that it should be determined to this

same accuracy which is about a factor of five better than the present

QHE result 18: RH = 6 453.2004 ft (± 0.17 ppm) for i = 4.

Resistors will continue to be used as working standards. But, if the

values of RH can be measured to sufficient accuracy and if for a given

sample they are shown to be time independent, then it will undoubtedly

be used to maintain national units of resistance, ft*, just as the Josephson voltage is used to maintain the volt; the quantized Hall effect

would then become the Josephson effect of the 1980's for electrical

metrology. (Note that the Josephson effect does not provide a useful

absolute voltage unit because the SI value of lelh is not yet known to

sufficient accuracy. The present uncertainty is about 2.6 ppm25

; this

uncertainty may soon be lowered to 0.1-0.5 ppm, but a change in the

accepted value of lelh of about 7 ppm26 can be expected.)

If the value of RH can be shown to be also sample independent and

to satisfy Eq. (5) to sufficient accuracy, then it can become an absolute

resistance standard. In order to verify that there is no unknown factor

involved in Eq. (5), the value of a should be independently determined

to higher accuracy. One of the attempts to do this is in progress at NBS

via a new yp experiment. 27 Absolute realization of the ft to the required

0.03 ppm level of accuracy via/?H will not be easy, but should ultimately

be possible.

SUMMARY AND UNANSWERED QUESTIONS

We have briefly reviewed the physics of inversion layer systems and

the discovery of the quantum Hall effect. If the quantum Hall effect is to be used for an improved determination of the fine structure constant

and the realization of a quantum standard of resistance, much more

information is needed about the experimental properties of inversion

layer devices, and further advances are required in our theoretical understanding of this unexpected phenomenon. The basic questions are: Why do Hall steps exist? Why are they so wide, flat and accurately

183
quantized despite imperfections in the devices? What are the fundamental theoretical and experimental limitations?

We have already seen how the special properties of two-dimensional

Landau level systems cause the transport to be dissipationless (p„ =

0) even in the presence of disorder. In Part II we shall explore the

implications of this important result and will examine progress toward

answering the questions listed above.

Acknowledgments

The work at NBS is supported in part by the U. S. Office of Naval Research and the Calibration Coordination Group of the Department of Defense and is being carried out

in collaboration with numerous colleagues from several institutions: R. F. Dziuba, B. F.

Field, A. C. Gossard, M. Jonson, C. F. Lavine, P. A. Lee, R. W. Rendell, P. J. Stiles, D. Syphers, N. N. Tadros, B. N. Taylor, D. C. Tsui and R. J. Wagner.

M. E. CAGE

Electrical Measurements and Standards Division,

National Bureau of Standards,

Washington, DC 20234

S. M. GIRVIN

Surface Science Division,

National Bureau of Standards,

Washington, DC 20234

References

1. K. von Klitzing, G. Dorda and M. Pepper, Phys. Rev. Lett. 45, 494 (1980).

2. T. Ando, A. B. Fowler and F. Stern, Rev. Mod. Phys. 54, 437 (1982).

3. A. B. Fowler, F. F. Fang, W. E. Howard and P. J. Stiles, Phys. Rev. Lett. 16, 901

(1966).

4. D. C. Tsui and A. C. Gossard, Appl. Phys. Lett. 38, 550 (1981).

5. Data from R. J. Wagner, C. F. Lavine, M. E. Cage, R. F. Dziuba and B. F. Field, Surf. Sci. 113, 10 (1982). For the data of von Klitzing see Ref. 1. See also S. Kawaji

and J. Wakabayashi, in Physics in High Magnetic Fields, Vol. 24, edited by S. Chikazumi and N. Miura, Springer Series in Solid State Sciences (Springer-Verlag,

Berlin, 1981), p. 284.

184
The Quantum Hall Effect II

The recent surprising observation of a quantization of the Hall resistance in units of

hie2 in quasi-two-dimensional conductors has necessitated a major rethinking of our

theoretical picture of transport in these systems. The central problem is understanding

why ideal behavior persists even in the presence of strong disorder.

INTRODUCTION

It has recently been discovered1 that the Hall resistance of the twodimensional electron gas is, under certain conditions, quantized in units

of hie2:

h

Rh = — (1)

An introduction to this striking phenomenon was presented in Part I of

this Comment. 2 In this second part we will discuss specific theoretical

questions of current interest. A consistent theoretical picture of this

phenomenon is only just beginning to emerge and there are many open

questions presently under active investigation.

THEORY

The quantum Hall effect is a vivid demonstration once again that physics

is an experimental science. The unexpected observation of this phenomenon has necessitated a major rethinking of our theoretical picture

Comments Solid State Phys.

1983, Vol 11, No. 2, pp. 47-58

0308-1 206/83/ 1 1 02-0047/$ 1 8 . 50/0

© 1983 Gordon and Breach

Science Publishers, Inc. Printed in the United States of America

185
of two-dimensional Landau level systems that is still in progress. We

reviewed in Part I the theory of transport in the idealized case of no

disorder and no Coulomb interactions. The central theoretical question

now facing us is Why do the Hall steps exist and why are they so broad,

flat and accurately quantized despite the presence of considerable ran- dom disorder in physically realizable devices?

We saw in Part I that the important feature controlling the transport

is the existence of a gap in the density of states at the Fermi level, or

at least a region of immobile states (a mobility gap) at the Fermi level.

At low temperatures all the states below the gap (i.e., below the Fermi

level) are filled and those above are empty. This means that there can

be no elastic scattering of the carriers and that the current flow is completely dissipationless (p^ = 0) even in the presence of impurities.

It is not at all clear, however, that the Hall resistivity (p^) should be

unaffected by the impurities. This is the question we now pursue.

In hindsight a hint of the existence of the quantum Hall effect actually

appeared in the theoretical literature even before the experimental discovery. Treating disorder in the self-consistent Born approximation

(SCBA), Ando et al? found that the Hall resistance is quantized when

the chemical potential is in the gap between Landau levels. However,

there should be essentially no experimental manifestation of this since

the chemical potential cannot be sustained for a finite range of carrier

density in a regime with zero density of states. When all the states

below the gap are filled, an infinitesimal increase in filling will cause

the chemical potential to jump discontinuously across the gap (since

states above the gap are now occupied). Thus the Hall step would have

zero width and the Hall resistance would be a smooth monotonic function of the magnetic field strength or gate voltage. In short, there would

be no quantization of the Hall resistance. In the limit of strong disorder

one expects no gap in the density of states—only a mobility gap due

to Anderson localization. This means that the region between the Landau

levels has a finite density of states but these states cannot take part in

the transport because they are localized, non-current-carrying states. The transport would still be dissipationless in this case and so could

produce a finite step width (as Ando et al? noted) but the SCBA is not

valid in this limit, and in any case no longer predicts quantization of

the Hall resistivity. Nevertheless, the quantization which was predicted

is indeed observed. 4

186
Having brought up the question of the plateau width let us deal with

it further. A second possible explanation of the width of the plateaus

does not require localized states to completely fill the gap between

Landau levels. Ando5 has suggested that if each Landau level is broadened due to some combination of smooth disorder and Coulomb interactions, there may be only a very narrow range of extended states near

the center of the level. This picture is consistent with recent discussions

within a percolation picture6,7 and could account for the extremely wide

Hall plateaus recently observed in GaAs heterostructures at very low

temperatures by Paalanen et a/.

8 The argument is that the Hall resistance

changes with carrier density only when extended states are being filled

and remains constant throughout the region of localized states. Another

possibility is that Coulomb interactions are so strong that some sort of

charge density wave9 (CDW) or crystallization occurs. Dissipationless

charge flow might then occur via some collective phenomenon. This

may produce the anomalously wide Hall steps, but our understanding

of this phase transition and its effect on the transport is rudimentary at present. A particular puzzle is that the nonlinear transport characteristics are unlike those of known CDW systems. There is no decrease in

resistance at higher currents as would be expected for depinning of a CDW. Rather the resistance (p^) increases with increasing current.

8

Still another mechanism, one which involves the tunneling of donor

electrons in heterostructures, has been put forth by Baraff and Tsui. 10

This may account for the increased step width in heterostructures relative

to MOSFETs.

Having discussed some of the questions associated with the width of

the plateaus, let us turn now to the accuracy of the quantization. Does

the existence of a mobility gap or a density of states gap imply quantization of the Hall resistance despite the presence of disorder? The first paper to address this question was presented by Prange 11 who solved

the case of a single delta-function impurity potential. This can be treated

exactly and he showed rigorously that even though the potential binds

an electron in a localized state, the Hall current is unaffected. The

current-carrying states are modified by the potential in such a way that

an extra current is produced which precisely compensates for the loss

of one of the charge carriers to the bound state. This result in the quantum

case is reminiscent of that for the classical two-dimensional system

discussed in Part I.

187
It seems reasonable that this current compensation can be expected

to hold for a finite density of impurities but a general proof is still lacking. Various authors 12-17 have put forth model calculations showing

quantization of the Hall resistance. These models have been very instructive, but all put some constraint on the form of the disorder or make some approximation, such as neglecting mixing between Landau

levels by the impurity potential.

One of the most interesting ideas concerning the disorder to be put

forth is found in the novel and very general approach taken by Laughlin. 1819

He argues that since the quantization appears to be so good experimentally, it must in fact be exact for a fundamental and simple reason,

namely, gauge in variance. Laughlin performs a gedanken experiment

in which the Hall current is carried in a ring so that periodic boundary

conditions apply. Consider threading a single quantum of flux through

the loop. One can show rigorously that the eigenstates and energy

eigenvalues are left unchanged by this operation—the states are simply

relabeled. It is important to realize that this is not actually a gauge

transformation. A gauge transformation modifies the vector potential in a way that produces no physically observable effects. In the present case the electrons feel a transient EMF since the inversion layer is acting

as a one-loop transformer secondary. This produces excitations and

thereby creates the Hall current. Thus Laughlin' s argument does not

actually invoke gauge invariance but rather takes advantage of a great

simplification which occurs when a single flux quantum is added. Because the eigenstates are precisely the same as they were before the

flux addition, the only change can be in the occupation of those states. Laughlin showed that if the flux is added slowly and if the electrons

respond to the perturbation adiabatically, then the net effect is to transfer an electron from one edge of the ring to the other. The relationship

between the induced current and the Hall voltage then automatically

satisfies the condition for quantization of the Hall resistance.

This beautiful argument has given us a good language in which to

discuss the transport. Further implications of the picture have been

explored by Imry. 20 Laughlin' s argument is, however, not completely

convincing to some.

21 One question centers on his claim that edge effects

pose no difficulty. 19 Let us analyze the situation beginning with the

adiabatic assumption. At first sight this would seem to be very bad

since we know that adiabatically increasing the flux through an ordinary

188
transformer (employing say, copper wire) leads to no current at all in

the secondary loop. This is because the secondary loop has finite re- sistance and the current disappears soon after the EMF transient. This

objection does not apply to the quantum Hall case: The mobility gap

at the Fermi level leads to dissipationless charge flow and so the current

should persist even after the EMF disappears. We can also give a well

defined meaning to adiabatic: The characteristic frequency w of the

perturbation must be such that hu> is much less than the energy width

of the mobility gap.

There is still a problem, however, because the gap must collapse at the edges of any real sample. Halperin22 has reformulated Laughlin's

argument to emphasize the fact that if is truly zero, then the current

produced in Laughlin's gedanken experiment will be the result of changes

in the current carried in the skipping orbits at the edges of the sample.

Halperin shows that ideal (impurity free) edges preserve the quantization

and argues (but does not prove) that, even in the presence of disorder,

the edge states should be extended current-carrying states which preserve

the total Hall current (within his particular model). It must be true,

however, that in real devices there is some dissipation at the edges and

so the current will not persist indefinitely. In addition, even if there is almost no dissipation at the edges, the lack of a gap implies that no

flux change can ever be truly adiabatic. How these points affect the

results of actual experiments is unclear at present.

FINITE SIZE EFFECTS

The question of edges brings up finite size effects in general. In a typical

Hall device there are only on the order of 109 Landau level electrons,

and 105 of these are in states at the edges. Since there are no errors observed at the level of accuracy, edge effects must be higher than

first order. Another finite size effect results from the peculiar pattern

of charge flow in the inversion region. Since is never truly zero,

the current cannot be strictly concentrated at the edges as in Halperin'

s

model discussed above. A standard macroscopic model23,24 has p^/p^

very large in the inversion layer and relatively small in the source and

drain contacts. Because the electrons flow along rather than perpendicular to the isopotential lines, the current entering and exiting the device

189
1

<I0

X4 o .£ '3 T3 o £P " § o- — a u

iiC £ <L» CO x: ,

l_

— 2 8? s > a s

=»

X o

V) (U p w)

N

CD o

(U 3

.5— «

<u 73 6

'•3 ii c o Oa « Q 22 *=a© ?

^53 O .C CC 00 g § o

*- "5 § O t/5 §** g.Oh 8 = •

_; o. ^ « & ° 2 % E <3 >

190
is concentrated almost entirely in diagonally opposite corners

23,24 as shown in Figure 1. This is a result of the Hall field being shorted in

the contacts so that the isopotential lines are forced to curve sharply near the ends. Model calculations23 - 24 have shown that this is not a

significant source of error in devices of aspect ratio greater than about

five. On the other hand this current crowding at the corners may produce

electron heating effects. The resulting lack of equilibrium could account

for the degradation of the plateau width which typically occurs for

currents above about 10 (jlA. Nothing definite about this phenomenon

is known at present.

There is one very important implication of this current crowding in

the corners. It leads to a "spraying resistance" for injecting electrons

into the device. Thus even though — 0 in the inversion region, there

is a large source-drain voltage drop. This is as it should be, of course,

since the applied source-drain voltage must exceed (or in this case

essentially equal) the induced Hall voltage. Unfortunately there is a

similar current crowding at the Hall probe contacts leading to an output

impedance for the device equal to the Hall resistance. This rather high

output impedance (—10 kfl) presents a major inconvenience to the

experimentalist attempting low noise, high sensitivity measurements.

COULOMB INTERACTIONS

So far we have talked about the effects of disorder and edges but have

largely neglected the role of Coulomb interactions. It is generally supposed that Coulomb effects will not destroy the mobility gap and so

the arguments given above will still hold, although again nothing rig- orous is known. One possible manifestation of the Coulomb interaction

may be found in cr^ when the chemical potential lies in the region of

extended states where 0. Paalanen et a/.

8 have observed a logarithmic temperature dependence of at very low temperatures (—100

mK) in GaAs devices. The logarithmic singularity may be due to en- hancement of Coulomb interaction effects at the Fermi level as a result

of the diffusive nature of the electronic motion in the presence of impurities. 25 - 26 These kinds of studies27 have proved useful in the "weak

localization" regime of metals and semiconductors and seem likely to

be important here as well. At small Landau level index (i < 4) Paalanen

et al.

8 see a suppression of far below the SCBA value, and also

191
observe an activated temperature dependence for o^. These effects are associated with anomalously wide Hall steps and, as mentioned earlier,

may be due to overwhelming Coulomb interaction effects which cause

a charge density wave (CDW) state to form.

Certainly the most intriguing recent event is the observation by Tsui

et al 28 of quantized Hall steps at RH = hle2

i with i = 3 and §. These

steps occur when the lowest spin state of the lowest level is 3 or § filled.

Related anomalies in cr^ have also been reported by Ebert et al.

29 The

leading candidate for an explanation of this anomalous quantum Hall

effect is a charge density wave formed at 3 and § filling. Laughlin has

speculated on the properties of this system. 30

Understanding the highly correlated ground state induced by Coulomb

interactions is one of the most difficult and exciting challenges facing us at present. The problem is that Hartree-Fock calculations show CDW

formation at all fillings with no preference for 3 or |. One supposes

therefore that the origin of the "commensuration energy" lies in some

Coulomb correlation effect neglected by the Hartree-Fock approximation. We believe that it is also fairly clear on physical grounds what

the nature of these correlation effects must be. There must be a fluctuation during which the constituents of a ring of three (or more) particles

cyclically trade positions. Such processes are familiar from Feynman's

roton theory of helium. 31 The quantum amplitude for such a fluctuation

event will contain a phase factor e

2lT,<t> where <|> is the number of flux

quanta in the area enclosed by the ring of particles. It is in this way

that information concerning the magnetic flux is transmitted to the electrons. The commensuration energy is thus dynamically generated by

these fluctuations. The smallest ring that encloses a nonzero area contains three electrons. Shifting each electron to its neighbor's position

involves two fermion exchanges giving a net plus sign. At a concentration of 3 the number of flux quanta contained in the electron crystal

unit cell is integral (three), so the associated flux phase factor is unity.

These points are presumably essential to the fact that 3 is favored.

One other observation from experiment is important at this point: The

anomalous Hall steps have finite width. That is, even when the concentration is not precisely 3 or §, the collective ground state still exists

and the correct quantization occurs. This suggests that the 3 and § fillings are sufficiently energetically favorable that the system prefers to form

domains of concentration 5 or § separated by domain walls which make

up for the excess (or deficit) of particles. Laughlin30 has speculated that

192
such domain walls may be essential to the transport and the quantization

at i = h and §. Developing a complete theory of the dynamic generation of the corn- mensuration energy will be difficult since it appears that no perturbation

expansion is possible: The kinetic energy has been quenched by the

magnetic field (the unperturbed Landau level is degenerate) and so the

Coulomb potential is not small in comparison to the kinetic energy.

One saving grace may be that one can probably neglect mixing between

the different Landau levels. This is because the observation of Hall

steps at both i = 3 and § suggests the existence of approximate particle-hole symmetry within the lowest Landau level. Such a symmetry

would not be present if there were significant mixing between different

levels.

THERMAL VOLTAGES

Another area of transport which is being explored is the thermoelectric

response of Landau level systems. It has been shown32 that on the

plateaus the thermopower in an ideal system is thermally activated and

so thermal voltages will (fortunately) be an exponentially small source of error in metrology experiments. However, when the chemical potential passes through the Mh Landau level, the thermopower has the

peculiar property of rising to a peak whose height is a universal number32

: S = In 2 (kB/e)/(N + \). Thermopower measurements offer us the

possibility of gaining new insight into the transport properties of inversion layer systems but unfortunately no data exist at present in this area.

RELATIVISTIC AND QED CORRECTIONS

Finally, we turn to the question of possible relativistic and quantum

electrodynamic (QED) corrections to the quantized Hall resistance. The

ratio of the magnetic energy hmc to the electron rest energy mec

2

is ~10"8

, which may eventually be a significant number relative to the

experimental precision. QED corrections received a limited investigation in connection with the Josephson effect and its metrological uses,

and similar questions need to be asked here.

193
First it should be noted that the Hall phenomenon is itself a relativistic

effect of first order in vie. One might suppose that there are higherorder effects arising from modification of the electron momentum-energy relation. However these cannot be important in comparison

to the severe modification of the momentum-energy relation due to the

semiconductor band structure. It is observed experimentally that even

the latter has no effect at least at the 10-6level of accuracy.

33 ' 34 Similarly, there may be QED effects arising from coupling to the photon

field. The measure of this coupling is the fine structure constant a =

e

2/hc. There is, however, another boson field present, namely the phonons, for which the coupling is vastly larger; very roughly, a ~ e

2/hvs where vs is the speed of sound. Hence vacuum fluctuation effects are negligible in comparison with possible solid state effects. We believe

that at zero temperature coupling to the fluctuating phonon field does

not destroy the mobility gap, and so one presumes that this effect poses

no difficulty, but to what level this persists is not known. Similar

considerations apply to vacuum polarization phenomena—the electron

gas is vastly more polarizable than the vacuum. In short, it would seem

that possible corrections should be sought in solid state effects and not

in anything more esoteric.

SUMMARY

In conclusion it has been found that the quantum Hall effect is significant

from the point of view of both metrology and fundamental physics. At

present it appears that all the metrology will be done on the Hall plateaus

where the Hall resistance seems to be universal, whereas all the new

physics and new information concerning transport will be obtained between the plateaus where sample and temperature dependent deviations

from ideal behavior occur.

Acknowledgments

The work at NBS is supported in part by the U.S. Office of Naval Research and the Calibration Coordination Group of the Department of Defense, and is being carried out

in collaboration with a number of colleagues from several institutions: R. F. Dziuba, B.

194
F. Field, A. C. Gossard, T. Jach, M. Jonson, C. F. Lavine, P. A. Lee, R. W. Rendell,

P. J. Stiles, D. Syphers, N. N. Tadros, B. N. Taylor, D. C. Tsui and R. J. Wagner.

S. M. GIRVIN

Surface Science Division,

National Bureau of Standards,

Washington, D.C. 20234

M. E. CAGE

Electrical Measurements and

Standards Division,

National Bureau of Standards,

Washington, D.C. 20234

References

1. K. von Klitzing, G. Dorda and M. Pepper, Phys. Rev. Lett. 45, 494 (1980).

2. M. E. Cage and S. M. Girvin, Comments Solid State Phys. 11, 1 (1983).

3. Tsuneya Ando, Yukio Matsumoto and Yasutada Uemura, J. Phys. Soc. Jpn. 39,

279 (1975).

4. The first observation of Hall steps appears to have been made by S. Kawaji and J. Wakabayashi in Physics in High Magnetic Fields, Springer Series in Solid State Sciences 24, edited by S. Chikazumi and N. Miura (Springer, Berlin, 1981), p. 284.

However, these authors did not demonstrate the accurate quantization of the Hall

resistance and their results did not appear in print until after those of Ref. 1

.

5. T. Ando, Surf. Sci., in press.

6. R. F. Kazarinov and Serge Luryi, Phys. Rev. B 25, 7626 (1982).

7. S. V. Iordansky, Solid State Commun. 43, 1 (1982).

8. M. A. Paalanen, D. C. Tsui and A. C. Gossard, Phys. Rev. B 25, 5566 (1982).

9. H. Fukuyama and P. M. Platzman, Phys. Rev. B 25, 2934 (1982).

10. G. A. Baraff and D. C. Tsui, Phys. Rev. B 24, 2274 (1981).

11. R. E. Prange, Phys. Rev. B 23, 4802 (1981).

12. H. Aoki and T. Ando, Solid State Commun. 38, 1079 (1981).

13. D. J. Thouless, J. Phys. C 14, 3475 (1981).

14. R. E. Prange and Robert Joynt, Phys. Rev. B 25, 2943 (1982).

15. T. Ando, Surf. Sci. 113, 182 (1982).

16. D. J. Thouless, M. Kohmoto, M. P. Nightingale and M. den Nijs, Phys. Rev. Lett. 49, 405 (1982).

17. P. Streda, J. Phys. C 15, L717 (1982).

18. R. B. Laughlin, Phys. Rev. B 23, 5632 (1981).

19. R. B. Laughlin, Surf. Sci. 113, 22 (1982).

20. J. Imry, J. Phys. C 15, L221 (1982).

21. J. J. Quinn and B. D. McCombe, Comments Solid State Phys. 10, 139 (1982).

195
22. B. I. Halperin, Phys. Rev. B 25, 2185 (1982).

23. S. Kawaji, Surf. Sci. 73, 46 (1978).

24. R. W. Rendell and S. M. Girvin, Phys. Rev. B 23, 6610 (1981).

25. A Houghton, J. R. Senna and S. C. Ying, Phys. Rev. B 25, 2196, 6468 (1982).

26. S. M. Girvin, M. Jonson and P. A. Lee, Phys. Rev. B 26, 1651 (1982).

27. B. L. Altshuler, A. G. Aronov and P. A. Lee, Phys. Rev. Lett. 44, 1288 (1980).

28. D. C. Tsui, H. L. Stormer and A. C. Gossard, Phys. Rev. Lett. 48, 1559 (1982).

29. G. Ebert, K. von Klitzing, C. Probst and K. Ploog, preprint.

30. R. B. Laughlin, preprints.

31. R. P. Feynman, Statistical Mechanics (Benjamin, Reading, Massachusetts, 1972).

32. S. M. Girvin and M. Jonson, J. Phys. C, submitted.

33. D. C. Tsui, A. C. Gossard, B. F. Field, M. E. Cage and R. F. Dziuba, Phys. Rev.

Lett. 48, 3 (1982).

34. K. Yoshihiro, J. Kinoshita, K. Inagaki, C. Yamanouchi, J. Moriyama and S. Kawaji,

J. Phys. Soc. Jpn. 51, L5 (1982).

196
Comite Consultatif d ' Electricite

e 16 Session 1983

PROGRESS REPORT

ON

INVESTIGATIONS OF THE QUANTUM HALL

EFFECT AS A POSSIBLE RESISTANCE STANDARD*

Contribution

from the

National Bureau of Standards

Washington, DC USA

This progress report to the Comite" Consultatif d ' Electrici te summarizes

our efforts to date (February 1983) to investigate the feasibility of using

the quantum Hall effect (QHE) as a new standard of resistance based solely on fundamental constants of nature. Succinctly stated, the results obtained

have been encouraging and indicate that the QHE could indeed be used as a resistance standard with parts in 10 8 precision . However, we believe that

the full inherent accuracy of the QHE cannot be realized until we overcome a puzzling problem that we have encountered during the course of the work. In

this report we present a chronological account of our QHE measurements and

our attempts to understand this problem.

It is now well known1 that the Hall resistance R^ of a two-dimensional electron gas is, under certain conditions, quantized in units of h/e2

:

where i is a small quantum integer, h is the Planck constant, e is the

elementary charge, y Q is the permeability of vacuum, c is the speed of light

in vacuum, and a is the fine-structure constant. These conditions can be

obtained by placing special, cryogenically cooled semiconductor devices in

large magnetic fields.

Rt,, defined as the ratio of the Hall voltage across the device to the

current through it, can be potentiometrically compared 2 with a calibrated

reference resistor Rr in series with the device. The relationship between

the Hall resistance and the series reference resistor is Rjj = ( v h/v

R^

* R

R' where is the voltage drop across Rn, provided the same current passes

through both the sample and the reference resistor.

* The work summarized in this report has been performed by M. E. Cage, R.

F. Dziuba, and B. F. Field of the Electrical Measurements and Standards

Division of the National Bureau of Standards in collaboration with these

colleagues from other institutions: R. J. Wagner (Naval Research

Laboratory), D. C. Tsui (Princeton University), A. C. Gossard (Bell

Laboratories), and N. N. Tadros (Guest Worker at NBS during 1982 from the

National Institute for Standards, Egypt). The NBS contribution to this work

is supported in part by the U.S. Office of Naval Research and the

Calibration Coordination Group of the U.S. Department of Defense. This is an informal report not intended for publication .

197
Our particular measurement approach is indicated in Fig. 1 . A constant

current is applied to a semiconductor device, cooled to 4.2 K or 1.5 K, and

to a room temperature reference resistor. Reference resistors have been

constructed with values nominally equal to that of the Hall resistance, i.e.,

6,453.2 0, for i=4 quantum Hall steps and 12,906.4 Q for i = 2 steps.

Therefore « Vr. The potentiometer voltage is made almost equal to the

voltage drop across Vjj or VR

, and an electronic detector D amplifies the

difference voltage signal. The isolated detector output is either displayed

on an X-Y recorder or is digitized for further processing by a desktop

computer. The current source, potentiometer, and electronic detector are all

battery-operated. Thermally induced emfs and linear drifts in the current

source and potentiometer are cancelled by reversing the current through the

device and the reference resistor. A series of reversals is made for each of

two measurements of which bracket in time one measurement of V^.

In January 1982 we published a value of in a Physical Review

Letter^ based upon measurements on three GaAs-Al

x

Ga-j__x

As heterojunction

devices. The measurements were made in July 1981 at Bell Telephone

Laboratories (BTL) and had an estimated one standard deviation total

uncertainty of 0.17 ppm. We then did a similar experiment with four silicon

MOSFET devices in September 1981 at the Naval Research Laboratory (NRL). The

total uncertainty was again presumably 0.17 ppm, but the September 1981 value

of Rfl was approximately 0.8 ppm larger than the July 1981 value. Figure 2 shows the results of our subsequent attempts to resolve this disagreement as well as the results of the two series of measurements just described and our first measurements done in May 1981 at BTL. These data represent twenty-one

separate measurements over an eighteen month time span on sixteen different

samples (seven GaAs devices and nine Si devices). Three of the GaAs

measurements involved i=2 Hall steps; all of the other measurements were on i=4 steps. These data were obtained at BTL, NRL, and NBS

Our battery-operated potentiometric measurement system initially had a 0.2 ppm one standard deviation random or type A uncertainty for measurements

with a 25 yA device current. The measurement system was improved until the

statistical uncertainty was reduced to 0.025 ppm prior to the October 1982

measurement at NBS. Therefore, six measurements (requiring a total of 66

minutes) now provide a 0.01 ppm statistical uncertainty of the mean. To

obtain the error bars assigned in Fig. 2, the appropriate type A uncertainty

was root-sum-squared (RSS) with one standard deviation estimates of the known

systematic or type B uncertainties, which were as follows: (1) 0.02 ppm due

to circuit leakage currents (based upon intercomparison of two resistors in

place of the sample and the reference resistor); (2) a 0.05 ppm reference

resistor temperature correction uncertainty until November 1981 when each

reference resistor was temperature controlled to sufficient precision to

reduce this uncertainty to a neglible level; (3) a 0.10 ppm uncertainty in the

step-up from fyjjjs (maintained via five 1 Q resistors) to the reference

resistor values; and (4) a 0.10 ppm transportation shift uncertainty for the

reference resistors when taken to BTL or NRL, unless the resistor value

changed by more than 0.2 ppm between the time that it left NBS and when it was returned (i.e., the before/after values differed by more than 0.2 ppm).

Whenever that occurred the most probable resistor value was assumed to be

either the before or the after value (depending on our best judgement as based on the calibration data), whereas the error bars were centered about

the average of the before/after resistor values, with a transportion

uncertainty of half the shift. (This is why the error bars of Fig. 2 are not always symetrically placed about what we consider to be the most probable

198
values.) Note that the data of Fig. 2 have been expressed in SI units, but

that the current 0.08 ppm estimated one standard deviation total uncertainty

in the NBS ohm (due in part to a possible drift since its last SI

realization 4 via the NBS calculable cross capacitor in 1974) was not included

in the error bars. This is because the same ft^s/ft correction was applied to

each measurement and we did not want to mask any unknown error sources by an increase in the error bars.

Our measurements began with a preliminary experiment at BTL in May 1981

(see the first two data points at the bottom of Fig. 2). All measurements

at BTL were performed using GaAs devices usually cooled to 4.2 K. The May 8

value has a large uncertanty because the 6,453.2 ft reference resistor 6.45J1

before/after values changed by 0.48 ppm. The May 14 value is also uncertain

because leakage currents in the sample holder (due to absorbed moisture)

contributed an additional 0.15 ppm uncertainty component to the measurement.

The correct value for this sample is likely to be higher. (That was indeed

the case when it was remeasured on July 15.) The sample holder was modified

by replacing the lead wires with Teflon-coated wire for the July experiment

at BTL which we now describe.

In July 1981 two measurements were made on i=2 steps using the

12,906.4 ft reference resistor 12.9J1, and one measurement on an i=4 step

using reference resistor 6.45J1. The i=4 step result is drawn with a dashed

line because the step did not appear to be flat on the chart recording. On

the same day, July 17, we replaced the GaAs device with two

12,906.4 ft resistors (12.9J1 and 12.9J2) connected in parallel and compared

them with resistor 6.45J1 . This comparison indicated a 0.38 ppm discrepancy

in the assigned resistor values, even though the before and after

calibrations at NBS of all three resistors implied that none of them had

significantly changed. Because 12.9J1 and 12.9J2 were also compared with

each other at BTL and this comparison agreed with their assigned

calibrations, it seemed more likely that resistor 6.45J1 had shifted by 0.38

ppm at BTL rather than both of the 12.9 resistors. Consequently, the

Physical Review Letter results from these data were based upon the

calibrations of the two 12.9 kft reference resistors and the July 17, 1981

result denoted by the symbol A in Fig. 2 was used in Ref. 3 - except that

a +0.3 ppm correction (to be discussed later) had not been made.

We next did an experiment at NRL in September 1981 using four different

Si MOSFET devices and i=4 steps. The before and after calibrations at NBS of

6.45J1 did not show any significant change, but those for 12.9J1 and 12.9J2

did. Thus the most reliable values were based upon the calibration of 6.45J1

and were about 0.8 ppm larger than those obtained at BTL. This might

possibly have been due to a difference between values of Ry for GaAs and Si

devices

.

To test this possibility, we performed experiments in January and

February 1982 at NRL in which we (1) compared two GaAs heterojunctions and

five Si MOSFETs with resistor 6.45J1, and (2) compared directly GaAs

heterojunctions with Si MOSFETs. The sample versus 6.45J1 comparisons were

all in excellent agreement except for those samples in which the Hall step

did not appear to be flat in chart recordings for one or both current

polarities. The resulting deviations from Rfl nominal can have either sign,

and this sign appears to be associated with both rhe slope of the step and

with the structure on the side of the step. It is not surprising that such

deviations exist for imperfect steps because Eq. (1) holds only for flat

199
steps (i.e., only when there is an actual mobility gap at the Fermi level).

Obviously, one would only use samples as resistance standards for which there was a region that appeared to be flat within the measurement precision.

The two direct GaAs-Si comparisons were done by mounting a GaAs device

and a Si device in series on the same sample holder (header) and directly

comparing their Hall voltages using a 10 or 20 uA device current. The GaAs

device was tilted, via a wedge, with respect to the magnetic field (i.e., the

plane of the device and the field were no longer perpendicular) such that

when the magnetic field was adjusted to the center of the GaAs i=4 step the

field was sufficiently large to obtain a good i=4 step for the Si device

(-13T). The Si MOSFET gate voltage was then adjusted to the center of the Si

i=4 step.

Unfortunately, the data changed character during the course of the

GaAs ( 5

) -Si ( 5 ) comparison. The Si(5) gate voltage for the i=4 step was abnormally low that day, and the magnitude and sign of the data shifts are consistent with the Si device re-equilibrating such that the device was no longer biased on the flat part of the Hall step. Using only the initial

data, we found Rh(4) for GaAs(5) to be (0.04 ± 0.09) ppm lower than for Si (5)

[type A uncertainty only since it was a direct comparison experiment] . We

also found the GaAs (6) Rjj value to be (0.06 ± 0.06) ppm lower than Si (9).

Hence there is no statistically significant difference in R^(4) between the

GaAs and Si devices. We therefore conclude that the 0.8 ppm difference

between the BTL and NRL measurements is not due to Si -GaAs device

differences. (There may be small non-zero differences in the above

comparisons because at 10 yA, 13 T, and 1.5 K the Si steps are much narrower

than the GaAs steps. One may very well have to cool Si devices to "0.5 K if

they are to be used as resistance standards at the part in 10® level of

accuracy.

)

Several other tests were also made at NRL in January and February 1982

to try to resolve the 0.8 ppm difference between the BTL and NRL

measurements. They were: (1) a verification that the value of reference

resistor 6.45J1 had not shifted. This was done by measuring R^(4) at NRL for

the GaAs(5) device (which had a very good Hall step), recalibrating 6.45J1 at

NBS (observing no change in its value), and then obtaining the same result

for Rj| with GaAs(5) at NRL; (2) a check that tilting the GaAs devices did not

introduce a change in R^. This was done by obtaining the same value for

GaAs (5) when it was also tilted "10 about an axis perpendicular to the Hall

field direction; (3) verifying that there was no difference in Rfl whether

samples were mounted either on magnetic or nonmagnetic headers; and (4)

checking that the magneto-resistance of the reference resistors was negligible. This was done by moving 6.45J1 closer to the magnet, thereby

increasing the magnetic field strength at the resistor by two orders of

magnitude

The laboratory at NRL was thought to be electrically noisy, and we speculated that this noise might couple into the nonlinear devices and

introduce a systematic error into the results. Thus we felt that in order to

resolve the difference between the NRL and BTL measurements, the experiment

had to be done in a quiet environment at NBS. To this end, an 8T

superconducting magnet system was installed in a shielded room and the first

experiment performed in September 1982 using a GaAs device. The result was

in excellent agreement with the values of R^ obtained at NRL.

200
To further investigate the question of electrical noise, a method was devised to detect noise at the sample and to display it on a spectrum

analyzer. We found significant noise at the sample, and therefore

constructed a new sample probe with two-stage filters on each lead to the

sample. These filters reduced the noise at the sample from all external

sources by more than a factor of 100 (to below our detection limit).

Additional measurements in October 1982 showed that this 100-fold noise

reduction caused no change in Rjj to within 0.02 ppm. It was also found that:

(1) the value of is independent of the magnetic field direction to within

0.01 ppm; (2) the value of Rjj is current independent in this sample over the

range 10-25 yA to within at least 0.03 ppm at 1.5 K; (3) the Hall step is

flat to within 0.02 ppm over a magnetic field range that is 3% of the central

value, as shown in Fig. 3; and (4) the value of Rjj when using reference

resistor 6.45J1 again agrees with those values obtained at NRL and that this

value can repeatedly be reproduced to within 0.02 ppm if the reference

resistor is not physically moved. All of these results are very encouraging.

They indicate that the quantum Hall effect could indeed be used as a resistance standard to maintain a unit of resistance, and that this standard

would probably be much more time-independent than wire-wound resistors. As

q a matter of fact, we believe that we were able to detect parts in 10

day-to-day variations in resistor 6.45J1

.

Even though many questions and doubts have been addressed by this work,

the 0.8 ppm difference still remains. It might be due to a difference in

values of Rjj between i = 2 and i=4 steps, but this possibility seems highly

unlikely because of the excellent agreement found by our Japanese colleagues

at ETL for the i=4, i=8, and i=12 steps in Si MOSFETs. 5 We will remeasure R H

for an i=2 step when a sample becomes available just to be certain that our value is not step-dependent.

Since our problem does not seem to be associated with the QHE samples, we are now concentrating our attention on possible problems in the

measurement and calibration of the reference resistors. The reference

resistors are calibrated at 1 mA with a room temperature current comparator

bridge, but are compared with the samples at 10-25 uA. To test the idea that

the resistor values might be current-dependent, the QHE measurement system

was used to compare resistor pairs over the current range 25 yA - 1 mA. We

did observe some changes in the resistor differences on the order of 0.1 ppm

when the current was varied from 25 yA to 1 mA. The changes seem to be

reproducible for some resistor pairs but not reproducible for other pairs.

Thus current-dependent effects could indeed be a problem for measurements

below the 0.1 ppm level of accuracy, but do not appear to be a major source

of our 0.8 ppm difference.

We also found that there was a (0.025 ± 0.005) ppm offset in our measurement system at 25 yA, and the same offset (with a larger uncertainty) at 1

mA. This offset was discovered by replacing the sample with a resistor,

measuring the resistor-reference resistor pair difference, and then

interchanging the two resistors in the circuit. Using this technique, we

then measured all six pair-wise combinations of three reference resistors

(6.45J1, 6.45E1, and 6.45E2) and corrected the results by the 0.025 ppm offset

We obtained loop-closure to within 0.03 ppm at both 25 yA and 1 mA.

Because the values of the three resistors differ by -30 ppm to -2 ppm from

the nominal value 6,453.2 Q, these loop-closure measurements check the

linearity of our QHE measurement system. However, systematic errors in the

system that are proportional to the resistor offset from nominal are not

201
detected by the loop-closure experiment. This system "gain" error was checked in another experiment in which the same three resistors (6.45J1,

6.45E1, and 6.45E2) were each compared with the GaAs ( 7 ) sample and with each

other. The results can be interpreted as a 2% "gain" error in the

measurement system. These results are very surprising because the null

detector used in the system (which would be primarily responsible for such an error) had been calibrated to a much higher accuracy. We are currently

investigating this problem. Extrapolating the results for 6.45E1 and 6.45E2

to zero nominal value yields values of that are in excellent agreement

with those shown in Fig. 2 for GaAs ( 7 ) using 6.45J1 (whose value is within

3 ppm of nominal )

Careful study of Fig. 2 indicates that the observed 0.8 ppm difference

could be correlated with the use of either a 6,453.2 ft resistor or a 12,906.4 ft resistor as the reference resistor. Indeed, when the two

12,906.4 ft resistors 12.9J1 and 12.9J2 are individually calibrated on the

current comparator bridge and then connected in parallel and calibrated as a

6, 453.2 ft resistor, there is a 0.3 ppm discrepancy between the measured value

of the parallel combination and the predicted value based on the individual

12.9 kft calibrations. (Measurements of the terminal resistances of 12.9J1

and 12.9J2 indicate that these should alter the predicted value of the

parallel combination by only 0.001 ppm.) All results shown in Fig. 2 that

involve parallel combinations of 12.9J1 and 12.9J2 (i.e., those denoted by

a 0 or by a A) have been corrected by 40. 3 ppm to agree with the actual

parallel calibration.

We plan to check the linearity of the current comparator bridge from

1 00 ft to 6,453.2 ft by constructing a Hamon device that consists of eight

800 ft resistors plus a 53.2 ft resistor. (Existing Hamon devices are used to

scale the ohm from the Thomas 1 ft resistors, which define the NBS ohm, to the

100 ft level, and thence to the 10,000 ft level.) An independent check is also

planned in which we will use purely resistance ratio measurements to scale

down from the 10,000 ft level to calibrate the 6,453.2 ft and 12,906.4 ft resistors

Conclusion

We are presently in the frustrating situation of being able to easily

reproduce values of that are in agreement for different samples to a precision of a few parts in 108

, but with an inaccuracy that may be as large

as one part in 1

6 . Our immediate goal is to reduce this inaccuracy to one part in 1 O^T Only then can we begin the tedious task of achieving the few

parts in 1

8 level of accuracy that we seek for an SI quantum Hall effect

resistance standard.

References

1. K. v. Klitzing, G. Dorda, and M. Pepper, Phys . Rev. Lett. 45_, 494

(1980)

2. M . E. Cage, R. F. Dziuba, B. F. Field, C. F. Lavine, and R. J.

Wagner, in Precision Measurement and Fundamental Constants II , Ed. by B. N.

Taylor and W. D~. Phillips, Natl. Bur. Stand. U.S., Spec. Publ. 617, to

appear in 1983.

3. D. C. Tsui, A. C. Gossard, B. F. Field, M. E. Cage, and R. F.

202
Dziuba, Phys. Rev. Lett 48, 3 (1982).

4. R. D. Cutkosky, IEEE Trans. Instrum. Meas. IM-23, 305 (1974).

5. K. Yoshihiro, J. Kinoshita, K. Inagaki , C. Yamanouchi, J. Moriyama,

and S. Kawaji, 16th Int. Conf . on Physics of Semiconductors, Montpellier,

France, Sept. 6-12, 1982.

6. L. Bliek, E. Braun, H. J. Englemann, H. Leontiew, F. Melchert, W.

Schlapp, B. Stahl, P. Warnecke, and G. Weimann, to be published in

PTB-Mitteilugen 93, (1983).

7. E. R. Williams and P. T. Olsen, Phys. Rev. Lett. £2, 1575 (1979).

8. R. 3. Van Dyck, Jr., P. B. Schwinberg, and H. G. Dehmelt, in Atomic

Physics 7, Ed. by D. Kleppner and F. M. Pipkin (Plenum, New York, 1981),

p. 340.

9. T. Kinoshita and W. B. Lindquist, Phys. Rev. Lett. _47_, 1573 (1 981).

Figure Captions

Figure 1 . A simplified schematic of the measurement circuit used to compare

the Hall resistance of a semiconducting device to a nominally equal room temperature reference resistor via a potentiometer adjusted to within a few

ppm of Vh« The entire system is battery-operated, including the electronic

detector, D.

Figure 2. A chronological ordering of precision measurements of R^ using

GaAs-Al^Ga^^As heterojunctions and Si MOSFET samples. These data are plotted as deviations A Ru i- n parts-per-million from the nominal value of

R^ (1) E25,312.80 ^ . Those data with points represented by the symbol X are

for i=4 steps compared with reference resistor 6.45J1; those with the

symbol ° are for i=2 steps compared with resistor 12.9J1; those with the

symbol • are for i=4 steps compared with resistors 12.9J1 and 12.9J2

connected in parallel; and those with the symbol A are derived results for

i=4 steps obtained by first comparing the sample with resistor 6.45J1 and

then adjusting that result by an amount indicated by a comparison of resistor

6.45J1 with the parallel combination of 12.9J1 and 12.9J2. Also shown are

the most recent values of R^ reported by ETL (Ref. 5) and by PTB (Ref. 6), as well as those based upon a yp(low) experiment (Ref. 7) and the

University of Washington anomalous magnetic moment of the electron experiment

(Ref. 8) plus QED calculations (Ref. 9). The measurements with dashed

error bars are questionable; see the appropriate comment below.

Comments

(a) This value of Rjj could actually be larger because of possible leakage

currents in the sample probe.

(b) The step did not appear to be flat in chart recordings, and hence this

value of Rfl is questionable.

(c) There was no before or after shift in the calibrated value of reference

resistor 6.45J1 for this measurement, and thus this is one of our most

reliable results.

203
(d) The gate voltage may have inadvertently been adjusted so that the sample

was slightly off of the step.

Figure 3. A high precision mapping of the i=4 step of sample GaAs(7),

obtained on October 21, 1982 at 25 yA and 1.5 K. The data points are plotted as deviations of Rfl(4) in ppm from the flat region of the step. The

larger random or type A uncertainty error bars are for data obtaned in 11 minutes while the smaller error bars are for data obtaned in 22 minutes. The

datum point with the open circle is for a different Hall potential probe set on the sample. The actual value of Rfl(4) agrees with that given in Fig. 2 for October 13, 1982 to within (0.005 ± 0.012) ppm.

204
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2/24/82 @ NRL

Si (9) i = 4

2/16/82 @ NRL

GaAs (6) i = 4

2/16/82 @ NRL

Si (8) i = 4

2/9 - 11/82 @ NRL

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2/8/82 @ NRL

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1/26/82 @ NRL

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1/22/82 @ NRL

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1/21 - 1/22/82 @ NRL

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9/17/81 @ NRL

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9/16/81 @ NRL

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207
IKEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-25, NO 4, DECEMBER 1976

Laboratory Voltage Standard Based on 2e/h

BRUCE F. FIELD, member, ieee, and VICTOR W. HESTERMAN

Abstract—The increased dependence of a standards laboratory on a stable voltage reference has made the maintenance of a "volt" by standard cells a costly and time consuming process. This paper

describes an instrument designed to calibrate cadmium-sulfate

standard cells directly against a time-invariant superconducting

Josephson junction voltage reference, thus replacing the large groups of cells typically used as a voltage reference. An induced

voltage of 5.2 mV is produced across the Josephson junction by ir- radiating the junction with microwaves of a known frequency. A

specially designed potentiometer is used to scale this voltage up to 1.01+ V where it can be compared to a standard cell. The overall accuracy (2 a) of the system is 0.4 ppm or better.

I. Introduction

APROTOTYPE voltage standard instrument

has been developed that is based on the ac Josephson effect in superconductors. Using a Josephson junction

as a precise time-invariant voltage reference is a well es- tablished technique which is used by many national lab- oratories [l]-[5]. The junction is used to produce a voltage

that is proportional to the frequency of applied microwave

radiation, where the proportionality constant 2e/h (e is the

electron charge, and h is Planck's constant) is independent

of individual junction characteristics. This instrument, similar (but less accurate) than the one

used to maintain the NBS Volt, is designed to be used by

a secondary standards laboratory to calibrate their stan- dard cells without being dependent on a national laboratory. The system consists of three major parts; the Jose- phson junction and dewar system, the junction sources, and the potentiometer. A description of each of these parts

follows.

II. Josephson Junction

The junctions used with the system are thin-film tunnel

junctions with an in-line geometry. The junction devices were developed at NBS by Finnegan [6]. The junction is composed of lead-lead -oxide-lead films deposited on a

glass substrate. The junction is coupled to a transmission

line formed between the film on the substrate and a ground

plane. This in turn is connected to 0.141" semirigid cable. The resonance frequency of the device is determined by

an external microstrip resonator deposited on the substrate

as shown in Fig. 1. This permits the resonance to be controlled more accurately during device fabrication. The

devices we are using have a resonance frequency of ap- proximately 9 GHz.

Manuscript received June 29, 1976. B. F. Field is with the Electricity Division, National Bureau of Standards, Washington, DC 20234.

V. W. Hesterman is with Superconducting Technology Inc., Mountain

View, CA.

Fig. 1. Josephson junction device geometry. A junction is located at each end of the resonator strip. (Only one is used for the measurements.)

The junction will be used as a fixed voltage reference operating at approximately 5.2 mV. Junctions with resis- tances of 0.5 to 2.0 have been characterized with current

step heights of 100 nA for the low-resistance junctions and

5 mA for the high resistance junctions. We have determined

that a current step height of 30 nA is necessary and we have

not as yet been able to operate the low resistance junctions

at 5.2 mV and 4.2 K. As it is very desirable to operate at 4.2 K, a compromise is needed between current step height

and junction resistance.

III. Dewar and Probe

The Dewar used with the system is a 30-1 liquid-helium

Dewar which is superinsulated and thus requires no liquid

nitrogen. It's liquid helium loss rate is less than 2 1/day.

This gives a 15 day hold time between refills. This dewar

contains three copper radiation shields that are cooled by

the cold helium gas venting out the 5-cm diameter neck

tube. The all metal design provides electromagnetic

shielding for the Josephson junction.

The Josephson junction, a superconducting shorting

switch and a ferromagnetic shield are mounted on the bottom end of a simple probe that is immersed in liquid

helium in the Dewar. The top end of the probe has a small SMA connector for the microwave, input, a knob to operate the superconducting shorting switch, a port to allow vacuum pumping

of the liquid helium to reduce the temperature, a small

pressure gauge to monitor Dewar pressure, a pair of copper

leads used for the output EMF of the Josephson junction,

and a connector for applying a bias current to the junction.

208
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1976

A 4.2 K mu-metal shield at the bottom of the probe

produces a low magnetic field for the junction. A ferro- magnetic shield could be used on the outside of the dewar

at 300 K but that would add considerable cost and weight.

An external shield could also be easily damaged by mechanically straining it. The superconducting switch consists of a pair of babbit

blocks separated by a fiberglass plate. A heavy niobum

banana plug can be pushed into a hole in the edge of fi- berglass plate thereby producing a superconducting short between the babbit blocks. The banana plug has sharp

edges that cut through surface oxide on the babbit blocks. The switch is actuated by a fiberglass rod that extends

through an "O" ring seal in the top of the probe.

IV. Microwave Source and DC Bias Supply

The microwave source to be used is a commercially

available source consisting of a phase-locked transistor

oscillator operating in L band and a frequency multiplier.

The output is approximately 9 GHz at up to 50 mW, which

is required to irradiate the junction. Additionally, an adjustable attenuator, a dc block (to eliminate ground loops),

and an isolator (to absorb reflected power) are placed in series in the output.

The high-stability frequency reference for the source is a temperature controlled 100-MHz crystal oscillator. This

oscillator has an initial long term stability of 0.015 ppm/

week. Due to the linear voltage to frequency relation of the

junction, an error in the frequency is reflected as a proportional error in the measured standard cell voltages.

Thus the frequency reference must be checked against

WWV (or a suitable frequency standard) and adjusted if necessary. The dc bias supply provides a stable dc current to keep

the junctions at the correct operating point, and contains

amplifiers with voltage offsets to permit observation of the IV characteristic of the junction on the oscilloscope. Observation of the junction IV characteristic is necessary to adjust the bias conditions to the proper operating point.

A rechargeable sealed lead-acid battery is used as the

source in the bias supply, so when measurements of the

junction voltage are being made the junction is independent of the ac power lines.

V. Potentiometer

The potentiometer designed for the system has an adjustable ratio of approximately 1/196. Instead of the usual procedure of "fine-tuning" the junction voltage to match

the potentiometer output, the potentiometer ratio can be

changed («0.3 percent) to accommodate both unsaturated and saturated standard cells. A simplified circuit diagram

is shown in Fig. 2. The design of the instrument is based on a method of

series and parallel connection of 14 nominally equal resistors. The ratio of the resistances (series connected/

parallel connected) is n

2

, and the error contributed by the mismatch of the resistors from nominal is second-order.

KELVIN VARLEY

CONSTANT

CURRENT

SOURCE

GALVANOMETER

Fig. 2. Simplified circuit diagram of the potentiometer, with the paralleling network for the main resistors omitted for clarity.

-15V

Fig. 3. Circuit diagram of the constant current source for the potenti- ometer.

The stepup from the junction voltage is accomplished as

follows. The 14 resistors are first connected in parallel.

Commercal grade rotary switches in series with compensating resistors (fans) are used to obtain a high accuracy

ratio. The 14 resistors are permanently connected together

using 4-terminal (tetrahedral) connections. The current

source in the instrument is adjusted to equate the voltage

produced across the parallel connected resistors, to the

junction voltage (at J - J'), using a photocell galvanometer

as a detector. Restoring the resistors to the series con- nected mode produces a calibrated voltage, across the se- ries-parallel resistors and the Kelvin-Varley divider, that can be compared to a standard cell. For ease of use it was decided not to require readjustment of the junction voltage and operating current of the instrument for each cell to be calibrated. Instead, a Kelvin-Varley divider shunted by a 13.786-J2 resistor is used

to provide a small adjustment of the nominal 1/196 ratio. Thus only periodic "standardization" of the current is needed to compensate for drifts in the current source. The

divider dials are arranged to be direct reading in volts with

a range of 1.0170 to 1.0199, and with a resolution of 0.1 ^V. Because of the limited adjustment range, only moderate

accuracy is required of the divider. The requirements for the constant current source are:

it should be unaffected by load changes, have good stability over a 2- or 3-h period, and have low noise output. The

209
FIELD AND HESTERMAN: LABORATORY VOLTAGE STANDARD

TABLE I Summary of Sources of Systematic Error in the Potentiometer

Sources Values

ppm (2o)

1. Main resistor mismatch <t>.01

2. Fan resistor mismatch <0.02

3- Resistance of tetrhedral Junctions <0.02

k. Main resistor self-heating *0.10*

5. Potentiometer temperature stability 0.01**

6. Kelvin-Varley divider and shunt 0.20

7. Current source stability 0.08

8. Leakage resistances <0.0*j

9. Effects of thermal EMP's 0.02*

•These values are subjective estimates intended

statistical 2 0 limits.

to be comparable with

circuit diagram of the current source used in the instru- ment is shown in Fig. 3. The reference for the source is three temperature regulated mercury batteries. The addition of a field-effect transistor at the output of the operational amplifier increases the effective dc loop gain of the system while preserving an adequate phase margin for

stability. The source exhibits drift on the order of 0.2 ppm/h which is due almost entirely to drift of the mercury

battery reference. Changes in the current due to changing

load resistances are less than 0.04 ppm for any setting of the Kelvin-Varley divider or series parallel connection of the resistors. Noise from the source is approximately 0.02 ppm rms in a 10-Hz bandwidth.

VI. Discussion of Errors

The estimated errors in a voltage measurement directly attributable to potentiometer uncertainties are listed in Table I and are discussed briefly below. High quality

wirewound resistors were used for the main series-parallel

resistors. These resistors had initial resistance matching

to better than 100 ppm and matching of temperature

coefficients of resistance to better than 2 ppm/°C. The

error due to resistance mismatch was measured after the

resistors were installed in the instrument. The self-heating error (more power is dissipated in the resistors in the series mode than in the parallel mode) was estimated from

measurements made on similar resistors. The error in the Kelvin-Varley divider and shunt

combination is a function of the dial setting of the divider.

If corrections were applied to the readings, the uncertainty

could be reduced to less than 0.01 ppm. However, it is anticipated that in routine operation corrections will not be

applied. For any dial setting the measured worst case error was less than 0.1 ppm.

VII. Results and Conclusions

A system has been developed to permit a standards

laboratory to maintain a unit of voltage without time

consuming standard cell calibrations in-house or at a national laboratory. The system uses a Josephson tunnel

junction as a time invariant voltage reference, and a special

potentiometer to scale the 5.2-m V output of the junction

to the voltage of a standard cell. An estimate of the uncertainty in a voltage measurement (comparable with

statistical 2a limits), which includes allowances for the potentiometer uncertainties and random measurement

uncertainties, would be 0.4 ppm. This uncertainty may be

reduced by a factor of three if corrections were determined

and applied to the Kelvin-Varley divider readings, and the galvanometer output was time averaged to reduce the random uncertainty. Difficulties in fabricating a junction

operable at 4.2 K with large step heights has prevented

final measurements from being made at the present

time.

Acknowledgment

The authors are grateful to L. B. Holdeman and T. F. Finnegan for numerous suggestions and help in the measurement of the junction characteristics. We would also

like to thank J. Toots for fabricating the junctions for the system.

References

[1] 1. K. Harvey, J. C. MacFarlane, and R. B. Frenkel, "Monitoring the NSL-standard of EMF using the ac Josephson effect," Metrologia,

vol. 8, pp. 114-124, 1972.

[2] J. C Gallop and B. W. Petley, "Recent NPL work on the AC jose- phson effect as a voltage standard," IEEE Trans. Instrum Meas.,

vol. IM-21, pp. ai 0-314, Nov. 1972.

[3] B. F. Field, T. F. Finnegan, and J. Toots, "Volt maintenance at NBS

via 2e/h: A new definition of the NBS volt," Metrologia, vol. 9, pp. 155-166, Mar. 1973.

[4] V. Rose, F. Melchert, W. Engelland, H. Fack, B. Furhrmann, P. Gutmann, and P. Warneck, "Maintaining the unit of voltage at PTB

via the ac Josephson effect," IEEE Trans. Instrum. Meas., vol. IM-23, pp. 271-275, Dec. 1974.

[5] G. H. Wood, A. F. Dunn, and L. A. Nadon, "The AC Josephson Effect Monitoring of the Unit of EMF in Canada," IEEE Trans. Instrum

Meas., vol. IM-23, pp. 271-278, Dec. 1974.

[6] T. F. Finnegan, J. Toots, L. B. Holdeman, S. Wahlsten, P. Mukhopadnyay, and T. Claeson, "Coupled Josephson tunnel junctions and

microstrip resonators," Bull. Amer. Phys. Soc, vol. 21, no. 3, p. 340, Mar. 1976.

210
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-23, NO. 4, DECEMBER 1974

Cryogenic Voltage Comparator

RONALD F. DZIUBA, member, ibbb, BRUCE F.

Abstract—The design and operation of a cryogenic voltage comparator system for precision 2e/h measurements is described. Major

improvements embodied in the new 2e/h system include the use of

1) a single microstripline-coupled Josephson tunnel junction to obtain usable step voltages up to 10 mV at 10.0 GHz, 2) a cryogenic

voltage divider comprised of two resistors whose ratio is calibrated with a low-temperature dc current comparator, 3) a superconducting

quantum interference device (SQUID) null detector, and 4) super- conducting switching. The accuracy of the present 196:1 divider system is estimated to be about 2 parts in 10s on the basis of pre- liminary tests and is limited by resistor self-heating during

calibration.

I. INTRODUCTION

IN RECENT YEARS the use of the ac Josephson

effect to measure 2e/h has provided the basis for a new maintenance voltage standard [1], [2j. This low- temperature solid-state physical phenomenon is currently

in use at the National Bureau of Standards (NBS) [3]

and other standards laboratories throughout the world. Improvements in present prototype 2e/h measurement

systems to make them more simple, reliable, and portable

will permit widespread use of these systems as local lab- oratory voltage standards and as portable volt transfer standards. The successful realization of rugged compact

microwave-coupled Josephson devices and all cryogenic

voltage measurements systems are generally regarded as

significant steps in this direction. In this paper, we describe a new cryogenic 2e/h system

being developed at NBS. In contrast to cryogenic systems

previously reported [4], [5] in which the voltage ratio

is established via the series-parallel technique, the present

system consists primarily of a cryogenic voltage divider comprised of two resistors whose ratio is calibrated with

a low-temperature dc current comparator and a single

Josephson tunnel junction coupled to external microwave

fields via a microstrip transmission line. Superconducting

switching and a superconducting quantum interference device (SQUID) null detector complete the major cryogenic portions of the system. The aim of this paper is to describe the design features and the measured performance

of this new 2e/h system. In Section II, the Josephson device, which provides the low level voltage for comparison with the 1 V of a standard

cell, is discussed. The voltage comparator system, which

is operable in two principal configurations (i.e., cali- bration and measurement), is described in Section III.

Manuscript received July 3, 1974; revised August 9, 1974. The authors are with the Electricity Division, National Bureau of Standards, Washington, D.C. 20234.

System for 2e//i Measurements

, MEMBER, IEEE, AND THOMAS F. FINNEGAN

The results of preliminary testing of a 196:1 voltage

divider are presented in Section IV.

II. JOSEPHSON DEVICES

A Josephson junction can be made in a variety of ways

by weakly coupling two superconductors, e.g., via a thinfilm oxide, a point contact, or a constriction in a film. When the junction is placed in an applied microwave

field of frequency v, a series of steps can be induced in the

current-voltage characteristic at voltages Vn = nhv/2e,

where n is an integer, and the junction can be regarded as

a precise frequency-to-voltage transducer. The maximum

usable value of Vn depends on the particular device; however, the tunnel junction has been applied the most

successfully in attaining large voltages, Vn > 5 mV. By

coupling microwave radiation to a series connection of resonant tunnel junctions via X-band waveguide, total Josephson voltages of 10 mV have been regularly obtained

and routinely used in 2e/h measurements both with inline [2] and cross-type [3] geometries.

Recently, a new method of coupling microwave radiation

to Josephson tunnel junction devices has been developed

in which the thin-film tunnel junction is mounted on a ground plane and forms an integral portion of a 50-Q

microstrip transmission line [6]. To be useful in a cryogenic 2e/h measurement system, the device should pro- vide a large voltage and the device holder should be compact and well shielded. The microstripline-coupling tech- nique has been refined and used to produce a Josephson

device specifically for the 2e/h application.

The device itself is composed of a single in-line Pb-Pb

oxide-Pb Josephson tunnel junction (1.0 by 0.3 mm)

deposited on a glass substrate. The narrow Pb strip (0.3 mm) is deposited first. After thermal oxidation of this

film, a second Pb film (2 mm wide) is deposited to form

both the junction and a 50-ft transmission line when the

substrate is mounted above a ground plane. Four gold

films which had been previously evaporated provide the

voltage monitor and "potentiometer" leads. The junction

is mounted in a special microwave package as shown in

Fig. 1. The microwave fields are coupled to the top of the

cryostat via semirigid 50-fl coaxial cable. The center con- ductor of the cable and the feedthrough at the end opposite

the microstripline launcher are used to feed a dc bias current to the junction. The voltage and potentiometer

leads are also connected to feedthroughs which consist

of lossy inductors containing iron epoxy filler for filtering. With devices of this type, usable junction voltages Vn between 5 and 10 mV were readily attained. The relevant 2e/h parameters for one of these junctions, resonant at

211
dziuba et al.: cryogenic voltage comparator

CONNECTOR TO MICROWAVE

TRANSMISSION LINE

POTENTIAL LEADS \

: —<tV

b-ryd

zr>

J

Fig. 1. Microstripline-coupled Josephson junction holder with top cover removed. The junction substrate (2..">4 by 2. 54 by 0.12 cm)

rests directly on a copper ground plane and microwave radiation

is coupled into the holder via semirigid cable.

SQUID READOUT

-V H

fl N R 30 U

JUNCTION

(a)

STD CEIL 3GALVANOMETER

TABLE I MlCROSTRIPLINE-CoUPLED 2e/k DEVICE PARAMETERS OF AN

In-Line 0.1-n Pb-Pb Oxide-Pb Junction (u = 10.0 GHz) at 1.9 K

V„ (mV) /. (aA) P (raW) Ratio

5.2 210 '20 196:1

7.1 170 35 144:1 10.2 100 75 100:1

Note: The corresponding fixed divider ratio required is also shown.

10.0 GHz, are shown in Table I. The maximum step amplitudes /„ are quite large even at 10 mV. (The external magnetic field is < 0.1 /iT.) The variation in the applied microwave power P be- tween 5 and 10 mV agrees well with that expected from

a simple analysis of the Bessel function dependence of the steps (i.e., P = kV2dc ) . This particular device was

cycled between cryogenic and room temperatures at

least six times 'with no noticeable change in its characteristics. These results make it clear that compact singlejunction devices can be achieved and are practical to use

in 2e/h measurements.

III. CRYOGENIC COMPARATOR SYSTEM

The cryogenic voltage divider consists essentially of two resistors and an external power supply. The divider

ratio N2

: 1 is calibrated with a low-temperature dc current comparator ( which establishes the ratio N: 1 ) in a twostep procedure by means of an additional auxiliary re- sistor as shown in Fig. 2(a). The choice of the nominal

divider ratio and hence the integer N depends on the ratio of the standard cell voltage to the usable Josephson junction output. The current comparator contains 17 ratio windings enclosed in a superconducting lead sheath wound

around a SQUID sensor and is designed to provide any

integer ratio up to 15:1. During calibration, each comparator—divider bridge circuit is balanced (with the SQUID null detector) by supplying a known error current

to an additional winding. For voltage comparison measurements with a Josephson

junction and standard cell, the cryogenic resistive divider

is connected to an external power supply to establish a

S0UID READOUT

n'r NR R T=2K

(b)

Fig. 2. Simplified circuit diagram of cryogenic voltage comparator

system, (a) Calibration mode, (b) Measurement mode.

voltage ratio (N2 :l) as indicated in Fig. 2(b). With a nominal 7-mV junction, for example, a ratio of 144:1 is suitable. In operation, the voltage ratio of the divider is fixed, and the null detector balances are made by adjusting

the current through the divider and the frequency of the

applied microwave radiation. By immersing the critical dc components in a (superfluid) liquid helium bath, the effects of thermal EMF's

are reduced, detector sensitivity is increased, and a stable uniform temperature is more easily achieved. The only

major parts of the apparatus not at cryogenic temperatures

are the microwave generation and frequency measurement

equipment, the SQUID readout system, and the power

sources.

A. Voltage Divider

In the prototype 2e/h system constructed for pre- liminary testing, a divider ratio of 196:1 was conser- vatively chosen to measure a voltage of 5.2 mV. The

main (two-terminal) resistors of the divider were construc- ted by loosely winding silicon-copper alloy wire [7]

(96-percent Cu, 3-percent Si, 1-percent Zn) on polytetrafluoroethylene formers and have nominal values of

7, 98, and 1372 G. The wire for the two lowest value

resistors has a diameter of 0.125 mm while that for a 1372-

212
IK EH TRANSACTIONS ON INSTRUM KNTATION AND MBASURKMKNT, DKCRMBKR 1974

Q resistor has a diameter of 0.025 mm (obtained by

drawing a portion of the 0.125-mm wire). Terminations

between the alloy wire and the superconducting wiring were accomplished via an intermediate copper terminal

directly hard-soldered to the resistance wire and are

similar to those described by Harvey and Collins [8]. The average fractional temperature coefficient (measured

between 2.0 and 4.2 K) of the 7- and 98-ft resistors was

-5.2 X 10-«/K and that of the 1372-G resistor, -6.2 X

10~*/K. Self-heating measurements indicated a load coefficient of -1 X 10-8/mW at 4.2 K and about -2 X

10-8/mW at 2 K. The divider power supply employs an unloaded mercury battery as the critical regulating element and is similar to that described by Finnegan and Denenstein [9]. The only essential difference in the present system is that

the main resistor (~300 ft) is located in the liquid helium

bath to minimize the effects of lead resistance variations caused by changes in the height of the liquid helium level

in the Dewar. An electronic dc temperature control system is used to regulate the temperature of the superfluid bath to < 1 mK

at any nominal temperature between 1.5 and 2.1 K. The resistance of a calibrated germanium thermometer

is monitored via a dc resistance bridge to determine the temperature, and the amplified bridge output is used to power a resistive heater (1.5-kft resistor) located in the

liquid helium.

B. DC Ratio Transformer

The current comparator used to calibrate the resistive divider is similar to that described elsewhere [10], [11].

Our comparator, designed to produce any precise integer

ratio up to 15:1, has been constructed by enclosing 17 insulated superconducting wires or ratio windings in a superconducting (lead) cable which was then wound

around a multihole SQUID sensor. An extra winding is included in order to introduce a small known current for balancing the SQUID detector during calibrations. The

number of ratio windings actually used depends on the

ratio of the particular resistive divider; e.g., in the pre- liminary tests described here, 16 of the 17 ratio windings were used to achieve a 14: 1 ratio. The SQUID device and cable containing the ratio windings were shielded with a lead-plated brass can.

All connections between and to the ratio windings were made some distance from the SQUID to eliminate end

effects. An upper limit on the current ratio accuracy of 2 X 10-9 was obtained by systematically checking 15 of the windings in succession against a reference winding at a current level of 150 mA. The unit winding current sensitivity of this comparator is 6 X 10-u A.

Calibration of the resistive divider is done in a twostep procedure with an auxiliary resistor as indicated in

Fig. 2(a). An inherent limitation of the present method is the self-heating in resistors R and NR, which are cali- brated with a current much larger (N times) than that used during measurements.

C. Null Detectors and Switching

The low level junction voltage balances are made with

the SQUID system of the current comparator. Its sensitivity in this mode of operation is 2.8 X 10_U V. The standard cell balances are made with a conventional

photocell-galvanometer amplifier whose isolated output is displayed on a recorder as in regular 2e/h measurements

[3].

'

All of the critical switching between the current comparator, resistors, power supplies, junction, and standard

cell for calibration and measurement is done with a rotary

switch. The switch which has five positions (and eight

poles) was rebuilt with fixed Babbitt metal contacts and

niobium-clad rotors. Each rotor consists of a phosphor

bronze spring with a small piece of niobium foil spotwelded to the contacts to insure a superconducting path

between the fixed Babbit contacts. (The contact force wras estimated to be 300 g.) The critical current of every

switch contact was tested and found to exceed 120 mA.

Repeated cycling between 2 and 300 K and over 600

contact closures did not appear to effect either the electri- cal or mechanical properties of the switch.

IV. RESULTS AND CONCLUSIONS

Although high precision 2e/h measurements have not

yet been made with the system, a preliminary evaluation

of its overall performance has been made by testing the

individual components and subsystems. At present the

limiting source of uncertainty in the system is the resistor heating during calibration which introduces a deviation of about 2 X 10~7 in the 196 : 1 ratio. This deviation can be

determined to about 2 X 10-8(1 standard deviation).

Changes in the alloy resistors, when repeatedly cycled

between liquid helium and room temperature, were usually

less than 10-5 so that trimming is not a serious problem.

Adequate shielding of the various comparator components,

particularly the SQUID and tunnel junction, from one

another is an essential prerequisite for successful operation

of the overall system. In conclusion, we have designed and built a cryogenic

2e/h comparator system with a 196:1 ratio. The overall accuracy with which this system might be used to deter- mine 2e/h is about 2 X 10-8 based on the results of our

preliminary tests. Also, the successful development of

single microstripline-coupled Josephson junction devices with voltage outputs near 10 mV now makes more attractive the application of alternative cryogenic divider tech- niques such as the series-parallel method for 2e/h work.

ACKNOWLEDGMENT

The authors wish to thank D. B. Sullivan, B. N. Taylor,

J. Toots, and J. Wilson for useful discussions and valuable assistance. REFERENCES

(1] B. N. Taylor, W. H. Parker, D. N. Langenberg, and A. Denenstein, "On the use of the ac Josephson effect to maintain standards of electromotive force," Metrologia, vol. 3, pp. 89-98, 1967.

[2] T. F. Finnegan, A. Denenstein, and D. N. Langenberg, "AC

213
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-23, NO. 4, DECEMBER 1974

Josephson effect determination of e/h: A standard of electro- chemical potential based on macroscopic quantum phase

coherence in superconductors," Phys. Rev. B, vol. 4, pp. 1487-

1522, 1971.

(3| B. F. Field, T. F. Finnegan, and J. Toots, "Volt maintenance

at NBS via 2e/h: A new definition of the NBS volt," Metrologia,

vol. 9, pp. 155-166, 1973.

[4] D. B. Sullivan, "Low temperature voltage divider and null- detector," Rev. Sci. Instr., vol. 43, pp. 499-505, 1972.

[5] J. C. Gallop and B. W. Petley, "High-precision measurement of 2e/h that uses crvogenic electronic components," Electron.

Lett., vol. 9, pp. 798-799, 1973.

[6] T. F. Finnegan, J. Wilson, and J. Toots, "Interactions in small systems of coupled Josephson junctions at microwave fre- quencies," Rev. Phys. Appl., vol. 9, pp. 199-205, 1974; also "Coupling between Josephson junctions and microstriplines,"

presented at the Applied Superconductivity Conf., Oakbrook,

III., Oct. 1974.

(7] D. B. Sullivan, "Resistance of a silicon bronze at low temperatures," Rev. Set. Instr., vol. 42, pp. 612-613, 1971.

(8| I. K. Harvey and H. C. Collins, "Precise resistance ratio meas- urements using a superconducting dc ratio transformer," Rev.

Sci. Instr., vol. 44, pp. 1700-1702, 1973.

[9] T. F. Finnegan and A. Denenstein, " High accuracy potenti- ometers for use with ten millivolt Josephson devices. I: Double

series-parallel exchange comparator," Rev. Sci. Instr., vol. 44, pp. 944-953, 1973.

(101 I. K. Harvev, "A precise cryogenic dc ratio transformer," Rev.

Sci. Instr., vol. 43, pp. 16261 1629, 1972.

| llj D. B. Sullivan and R. F. Dziuba, "Low temperature direct current comparators," Rev. Sci. Instr., vol. 45, pp. 517-519,

1974.

Reprinted by permission from IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT

Vol. IM-23, No. 4, December 1974, pp. 264-267 Copyright 1975 by the Institute of Electrical and Electronics Engineers, Inc. PRINTED IN THE U.S.A.

214
Volt Maintenance at NBS via 2ejh: A New Definition of the NBS Volt*

B. F. Field, T. F. Finnegan, and J. Toots

Institute for Basic Standards, National Bureau of Standards, Washington, D.C. 20234, U.S.A.

Received: March 16, 1973

Abstract

This paper describes in detail the procedures, methods and

measurements used to establish a new definition of the U.S.

legal volt via the ac Josephson effect. This new definition has been made possible by the use of thin film tunnel junctions (capable of producing 10 mV outputs) and high accuracy

voltage comparators. The Josephson junction is used as a

precise frequency-to-voltage converter with a conversion

factor equal to 2ejh. A series of measurements of 2e/h has been

carried out at NBS referenced to the as-maintained unit of emf based on a large group of standard cells. Measurements made at reg\ilar intervals over a one year period (1971 to 1972)

indicate that the mean emf of this group of standard cells has

decreased about 4 parts in 10'. Primarily to remove the effects of this drift, on July 1, 1972 a new as-maintained unit was

defined by choosing a value of 2e/h consistent with the existing unit of emf. The adopted value of 2e/h is 483593.420 GHz/ Fnbs. The precision (one standard deviation) with which the new

unit of emf can be maintained with the present techniques and apparatus is about 2 parts in 10s

. The accuracy of the

present system is estimated to be 4 parts in 108

. Comparisons

of 2e/h systems at different national laboratories have been

limited by uncertainties associated with the physical transfer of standard cells. In order to determine the relative agreement

of the various 2e/h systems with precision better than 1 or

2 parts in 107, it appears desirable to compare 2e/h systems

directly by transporting one of them.

1. Introduction The ac Josephson effect has made possible the

realization of a new voltage maintenance standard at the National Bureau of Standards. Critical to this realization is the role played by a Josephson junction

which may be regarded here as a frequency-to-voltage

converter, where the frequency-to-voltage ratio is precisely equal to the combination of physical constants 2ejh (e is the electron charge and h is Planck's

constant). A variety of experimental tests (for material dependence, temperature dependence etc.) [1 —5] and theoretical investigations [6—9] of the Josephson

relation have been made which indicate that for ordinary Josephson devices (particularly tunnel junctions) with conventional current-voltage lead configurations the ratio is exact to at least a few parts

in 108. However, further experimental investigation

of this question appears desirable. Since the first high precision measurements of

2e/A by Parker, Taylor and Langenberg [10], the

potential significance of a precision voltage standard

based on 2e/h has been recognized throughout the world [1, 2, 11—14]. It was first demonstrated by

Finnegan, Denenstein, and Langenberg (FDL) that standard cells could be maintained with a precision of 10-7 via a Josephson 2ejh apparatus over extended

periods of time (5 months) [1]. Establishing a unit of

* Contribution of the National Bureau of Standards. Not

subject to copyright.

emf based on a fundamental physical phenomenon

such as the ac Josephson effect removes many of the inherent difficulties in using a large group of standard

cells to maintain a unit of emf. One of the most im- portant of these difficulties is the gradual long term

drift characteristic of the emf's of standard cells. In this paper we present a detailed account of the experiments involved in establishing a working Josephson-effect voltage maintenance standard at NBS.

Much of the apparatus and procedures are essentially

identical with those described by FDL [1]. Section II contains a description of the Josephson devices and

their performance. In Section III, we briefly describe the microwave and Josephson-device bias equipment.

Section IV contains descriptions of the two dc voltage

comparators including estimates of their respective

accuracies. The experimental procedures used to compare a standard cell with the Josephson voltage

are described in Section V. This section also contains

2e/h data for a local group of standard cells. In Section VI, the history of the NBS voltage maintenance

and dissemination program based on standard cells is briefly reviewed. Section VII contains the NBS 2e/h

results including a discussion of the overall experimental uncertainties. A comparison of recent (1971

to 1972) values of 2ejh is presented in Section VIII.

Finally in Section IX, the procedures used for the NBS voltage maintenance standard based on 2e/h are outlined and its current precision noted.

2. Josephson Device Fabrication and Performance

All of the 2e/h measurements made at NBS have

utilized resonant thin-film tunnel junctions. These

junctions are more difficult to manufacture than

point-contact junctions, and couple to the microwave

power only over a limited frequency range; however,

they can be used to achieve higher step voltages. The

ability to use higher dc voltages is a significant ad- vantage since it minimizes the effect of thermal emf's generated in the cryostat and the junction voltage

measuring leads, and requires less sensitivity in the

voltage null detector. Furthermore, thin-film junctions are permanent structures which require no

mechanical adjustment before each experiment nor

periodic cleaning and reconditioning.

The junctions we have used are of cross-type

geometry with four junctions deposited on a single 2.54 cm square glass substrate, as shown in Fig. 1. The dimensions of the junctions are approximately

1 mm by 0.3 mm with the films approximately 150 run

(1500 A) thick. This geometric structure has a reso- nance frequency at about 9 GHz. The junctions used

215
for the 2<?/A measurements have been of Ph-Pb oxidePl> construction. Some Sn-Sn oxide-Sn junctions have

been made but usuable steps at the voltage required

by our measuring instruments are not easily attained. The junctions are fabricated with a thermal eva- poration system. The first layer of Pb is evaporated on each glass substrate and then oxidized in 1/2 atm

of oxygen at a temperature of about 40 °C. (The substrates are heated by radiation from infrared heat lamps external to the vacuum chamber.) The second

layer of Pb is evaporated and the junction is over- coated with a layer of photoresist after removal from

the evaporator. The junctions are then stored in liquid nitrogen dewars at 77 K.

Shortly after fabrication the dc /- V characteristic and normal-state resistance of the junctions are measured. Junctions with nearly identical I-V characteristics, however, can differ greatly in frequency

response, coupling properties, and the rate of decrease

of the maximum step height (current amplitude) with

increasing dc voltage. Measurements of the dc junction

0 5 10mm

1 i 1

Fig. 1 Josephson-device geometry. The cross-hatched region

indicates the oxidized film which is partially covered by the second film to produce four junctions

characteristics are useful in detecting gross defects such as shorts bridging the oxide barrier which can markedly reduce the coupling to external microwave

fields. The normal resistance of the junction is, in theory, inversely proportional to the zero voltage

current which in turn is related to the current amplitude of a given step. Therefore the use of low resist- ance junctions should provide higher step amplitudes.

Unfortunately, in going to lower resistance junctions

(below 1/3 ohm) the maximum zero voltage current, we observe, increases monotonically, but becomes a

significantly smaller fraction of the theoretical limit. Typical junctions with resistances of 0.1 and 0.05 ohm

have maximum dc currents of 9 and 12 mA compared

to the respective theoretical limits of approximately

20 and 40 mA. Some of this reduction may be attrib- uted to the asymmetric self-field generated by the dc

bias current flowing through the junction. In selecting

junctions for voltage measurement, one must compromise on the choice of junction resistance; it should

be small enough to yield a reasonable zero voltage current and consequently large step amplitudes, and

yet be large enough to provide sufficient coupling for operation at reasonable step voltages and applied

microwave powers. We have found that the use of junction resistances near 0.1 ohm is a good choice. When a tunnel junction is irradiated with micro- waves, at frequency v, the 1- V characteristic shown

in Fig. 2 ( a), is modified so that steps appear at discrete voltage intervals given by the relation 2e Vn = nhv. The height of these steps decreases with increasing

junction voltage (Vn ). As the step amplitude get

smaller it becomes progressively more difficult to

jg. 2. Typical J- 1 characteristic of a Pb-Pb oxide-Pb tunnel

unction, a) With no applied microwaves. Vertical scale: 10 mA/div.; horizontal scale: 1 mV/div. b) With applied micro- waves; constant voltage steps induced at 5 mV. Vertical scale: 20<)fiA/div.; horizontal scale: *20(iV/div.

remain biased on a step, due to drifts in the dc bias current and microwave power level. For our present

apparatus steps smaller than 50 fxA are considered unusuable. Our junctions respond to microwave

power only near their resonance frequencies. Continuous adjustability of the junction voltages is made

possible by the high step numbers used, despite the

limited frequency range. The details of the coupling

between external microwave fields and our resonant

tunnel junctions are not yet clearly understood.

Qualitatively better coupling is observed for higher

resistance junctions.

216
In order to get usable step amplitudes at 10 mV we

have chosen to operate two junctions in series. For

reasonable operating conditions we can tolerate a few

tenths of a percent mismatch in the two resonance

frequencies. However, a half percent mismatch often means that the junction combination is marginal.

When using junctions connected in series in the same

microwave field, we have found it advantageous to cascade junctions of nearly equal resistance. If the induced microwave steps have a finite slope

(i.e. d7/dF^oo), the Josephson voltage is no longer

well denned. The slope of the steps is known to be

sensitive to external electromagnetic noise [15, 16] and extensive precautions have been taken (shielded room and equipment) to reduce this noise. The slope

is regularly checked by slowly varying the bias current up and down the step and looking for a voltage deflection on the null detector used with the measuring

instrument. We have never observed any finite step

slope to within the resolution of our null detector

(a few parts in 108

).

3. Microwave and dc Bias Apparatus In order to obtain an induced step at a desired

voltage, sufficient microwave power at a well-defined frequency must be coupled into the junction and a

stable dc bias current passed through it. a) Mircowave Equipment

The microwave apparatus, as indicated in the block diagram in Fig. 3, includes a klystron tube mounted in an oil bath (with air cooling) for micro- wave stability. The klystron is phase-locked to a continuously-adjustable quartz-crystal oscillator for frequency stability. The short term frequency stability (15 min) is about 1 part in 109

. The frequency is measured by a frequency counter with an internal converter and a resolution of 1 part in 109

. The iso- lator, shown in the diagram adjoining the main

attenuator, prevents unwanted reflection of micro- wave power into the klystron. The microwave generation equipment and the cryostat containing the

junction holder aro grounded separately, and a dc

block is placed in the waveguide to prevent circulating currents. The accuracy of the counter time base is regularly

checked against the U.S. frequency standard. A 100 kHz high stability oscillator is compared to the signal

from WWVB using a VLF phase comparator. This

oscillator, which is maintained within a few parts

in 10 10 of WWVB, is simultaneously used to check the counter time base. The drift in the counter time base

is predictable and changes 2 parts in 1010 /day.

b ) Junction Holder Description

The glass substrate on which the junctions are evaporated is mounted in the center of a piece of copper X-band waveguide. A slot is milled halfway

through the waveguide and the substrate rests in this

slot. A close fitting cover holds the substrate in place

and provides continuous guide walls. The leads to the

junctions are copper magnet wire soldered to the

junctions using indium solder. The bias lead wires pass through small grooves in the substrate cover to Teflon

standoffs outside the waveguide where they are sol- dered to leads from a vacuum tight electrical con- nector mounted in the top of the holder. The voltage

leads are from a single spool of copper magnet wire and are kept in close thermal contact by running them

to the voltage measuring instrument in a single piece of Teflon tubing. In the cryostat they are enclosed in a length of copper tubing that always crosses the helium level. From the copper tubing to the top of the

cryostat the leads are enclosed in a thin-wall stainless

steel tube. This arrangement minimized heat transfer and tends to keep the large temperature gradient at one location on the leads.

COMPARATOR - ATTENUATOR CRYSTAL DETECTOR

POWER

SUPPLY

STABILIZER ATTENUATOR

JOSEPHSON

DEVICE

Fig. 3. Block diagram of microwave generation and frequency measuring equipment

217
In order to maximize the microwave electric field at the junction a shorting plug is placed in the waveguide about 4 cm below the substrate. Since placement

of this short is extremely critical, it can be adjusted

from the top of the cryostat while observing the step

pattern on an oscilloscope. An iris placed above the

substrate to form a cavity as described by FDL [1] was not used for our measurements.

c ) Bias Circuit The two junctions used for the 2ejh measurements

are operated with independent bias current sources, similar to the ones described by FDL [1]. The indi- vidual supplies consist of two 14 ampere-hour mercury

cells connected in parallel, and adjustable resistors connected in series with the junction to adjust the

current. The current is reversible and has separate

fine adjustment controls for the forward and reverse current so that small adjustments can be made in the

bias current while in one direction without disturbing the settings for the other polarity. An /- V display on

an X-Y oscilloscope with differential inputs is provided

for each junction by monitoring the voltage across a

resistor in the bias circuit and the voltage developed

across each individual junction. Adjustable positional

offsets in the display circuit are used to suppress the

origin of the /- V curve to allow expansion of any

portion of the curve for viewing. The offset for the

junction voltage is calibrated so the proper step number

can easily be determined.

The bias circuit normally supplies dc for use when

measuring the junction voltage ; however, for viewing

the /- V characteristic an ac sweep is extremely useful. Two types of sweep circuits were constructed for use with our bias supply. A separate specially shielded transformer (connected to the 60 Hz power line) can

be connected in place of the batteries to provide an

adjustable ac sweep which is useful for observing the

I- V characteristic with no microwaves applied and

determining the critical current. The transformer is disconnected from the bias supply whenever measure- ments are to be made and a second incremental sweep

circuit is used. The second circuit is battery operated

and built into the bias supply. This sweep is super- imposed on a dc level and is used to observe a small portion of the characteristic. The sweep consists of an oscillator with an active filter to remove all but the fundamental sine wave. During voltage balance both

ac sweep circuits and the oscilloscope are completely

disconnected from the dc circuitry s6 that the supply

is isolated from grounded equipment with an insula- tion resistance greater than 1011 ohms.

4. Voltage Comparators

Two voltage comparators are presently in use at NBS. Both of these instruments provide a fixed 100

to 1 ratio and rely on the fact that the junction voltage

is adjustable to exactly 1/100 of the standard cell voltage. These comparators were built and first used

at the University of Pennsylvania and are described

in greater detail in reference [1]. The use of a fixed 100/1 ratio comparator reduces

the measurement uncertainty by eliminating the need

for an adjustable resistance network since the junction voltage can be varied to achieve a null balance. Both instruments can be self-calibrated in less than

an hour using no external resistance standards. The

use of two instruments operating on different principles with different major sources of systematic

error is desirable since it allows us to perform a final check on unsuspected sources of systematic error. The

two instruments were periodically compared and

were always found to agree within 3+3 parts in 108

, well within the estimated total instrument uncertainty

of 4 parts in 108

.

a) Series-Parallel Comparator

The design of the first voltage comparator (the one used for the 2ejh runs) is based on a method of double series-parallel exchange. If a group of n nominally equal resistors are connected first in series and then in parallel, the ratio of the resistances of the two combinations is n

2 with an error which is second

order in the resistor mismatch. Figure 4 is a simplified

circuit diagram of the comparator (SPC) showing the two sets of ten matched resistors, one set in series and

one in parallel. If the two sets of resistors are ex- changed (i.e., the set originally in series is reconnected

in parallel, and the other set reconnected in series) and

a second set of balances made, the effect of inequality of the total resistance of the two sets of resistors is reduced to second order by averaging the results of the two balances. Tetrahedral junctions and compensating resistors

(fans) for paralleling the network permit the use of commercial grade rotary switches despite their high

contact resistance, while maintaining a high accuracy

series-parallel transfer. A power supply using mercury

batteries and regulated by a mercury battery under

essentially no load, supplies stabilized current (1/2 ppm/h) to the resistance networks. The effects of

thermoelectric voltages in the circuit are eliminated by reversing the current in the instrument, the stand- ard cell, and the junction bias current. Although the

100/1 ratio can be determined to greater accuracy than that with which the individual resistors can be compared, the power dissipated in the networks

changes by a factor of 100 when they are switched

from series to parallel. To reduce the error resulting

from self-heating the resistors are mounted in a thick walled aluminum can filled with oil. The entire in- strument is enclosed in a temperature regulated air enclosure. The instrument sources of uncertainty in a 2e/h measurement for the past year are essentially the same as reported by FDL [1] as the instrument was

modified only slightly for use at NBS [12]. These

estimated uncertainties are listed in Table 1 and are described briefly below.

a) and b) Checks on the matching of the main and

fan resistors were carried out regularly. The uncertainties were estimated from the check data. In the

case of the main resistor matching, the second-order

corrections to the resistance ratio were calculated. The

correction (as a result of periodic trimming of the

resistors) was always negligible ( < 1 part in 109

). The

deviations for the fan resistors were also measured

and found negligible. The uncertainties attributed to these two errors are 0.4 and 1 parts in 108 respectively.

c) The transfer resistance error due to tetrahedral junction asymmetries introduced a measured uncertainty of 4 parts in 10*.

218
+ I mA

POWER SUPPLY

1 + STANDARD CELL

.018 V )

BIAS SUPPLY

Fig. 4. Simplified circuit diagram of series-parallel voltage comparator

Table 1. Sources of uncertainty in 2e/h associated with the

series-parallel instrument

Uncertainty parts

in 10s (one standard

deviation estimates)

a) Main resistor mismatch 0.4

b) Fan resistor mismatch 1

c) Resistance of tetrahedral junctions 0.4

d) Main resistor heating effects 2

e) Comparator temperature stability 0.3

f) Working current stability 0.2

g) Calibrating signal accuracy 1 h) Leakage resistances 1

i) Dielectric polarization

j) Effects of thermal emf's

0.2

0.5 RSS total 2.8

d) The self- heating effects for the main resistors were measured in situ using the bridge within a bridge

technique [17]. Using this procedure a correction of

3 parts in 108 was applied to the data with an uncertainty estimated to be 2 parts in 108

. 1

e) Since the series-parallel resistance ratio is not

established simultaneously, any change in resistance between the measurements will cause a first order

correction. The largest cause of this change is the change in internal temperature of the instrument. The

temperature regulated oven changes slightly as ambient temperature changes and thus introduces an

uncertainty of 3 parts in 109

. f) For the 2ejh measurements made before De- cember 1971, a standard cell balance was made after the junction balance and a correction applied for the

drift of the power supply. Since December we have

1 The instrument is to be modified by replacing the main

resistors with better matched, lower temperature coefficient

resistors to significantly reduce the error due to self-heating. In addition the fan resistors have been replaced and the temperature control improved.

begun regularly making standard cell balances both

immediately before and after the junction balance and linearily interpolating the results to the time of

the junction balance. The uncertainty contributed by

non-linear drift in the power supply is estimated to be

2 parts in 10*. g) The balancing procedure requires that a cali- brating signal be introduced to normalize the null- detector deflections. The uncertainty in calculating

this signal is mainly due to resistor aging and results

in a measurement uncertainty of 1 part in 108

. h) The effects of leakage resistance are estimated

by measuring the individual leakage paths directly.

The estimated uncertainty from this source is 1 part

in 108.

i) Small leakage currents are induced in insulators due to a component of the insulator dielectric polarization, and polarization currents are induced due to

piezoelectricity caused by mechanical stress on the

insulators. The polarization currents were measured

using an electrometer and contribute an uncertainty

of 2 parts in 109

.

j) The uncertainty due to the effects of thermal

emf's is caused by variations of the thermal emf's in the instrument and non- linear changes in the thermal

emf's in the leads from the junction to the instrument. Five parts in 109 is estimated for this uncertainty.

The root-sum-square total uncertainty is 2.8 parts

in 108 and represents the total uncertainty associated with the series-parallel system exclusive of the random

uncertainty of a 2ejh measurement.

b) Cascaded- Interchange Comparator

The second instrument develops a fixed 100 to 1 ratio by use of a voltbox optimized for self-calibration. A simplified circuit diagram of the cascaded-inter- change comparator (CIC) is shown in Fig. 5. The cali- bration of the voltbox is accomplished by using a

second identical voltbox as shown in the figure. With

219
VOLTBOX A VOLTBOX B

POWER SUPPLY

j ilmA

POWER SUPPLY

Fig. 5. Simplified circuit diagram of cascaded-interchange voltage comparator shown in the calibrate mode The voltage devel- oped across any one of the four 10 ohm resistors is 1/100 of the total voltage across the voltbox

Table 2. Sources of uncertainty in 2e/h associated with the cascaded-interchange comparator

Uncertainty parts

in 108 (one standard

deviation estimates)

a) Random uncertainty of calibration 2 measurement

b) Switch and power supply variations 2 during calibration

c) Trimmer lead resistance error 0.7

d) Comparator temperature stability 0.3

e) Working current stability 1 f ) Calibrating signal accuracy 0.5

g) Leakage resistances 0.4

h) Dielectric polarization 0.2

i) Effects of thermal emf's 0.5 RSS total 3.2

the switsches set as shown, a 10 ohm equal-arm

Wheatstone bridge is formed and the resistors are trimmed until a balance condition is achieved for both

positions of the interchange switch. The next position of the "select-ratio" switch then forms a 20 ohm

equal arm bridge using the two 20 ohm resistors and

the four previously equalized 10 ohm resistors. Cali- bration is continued in this manner until both boxes

are adjusted to a 100 to 1 ratio. By independently

powering the voltboxes, the usual need for lead compensation is eliminated. The advantage of this method

is that the resistors always operate at the same power

so that the self-heating is negligible. However, the

calibration procedure must be performed carefully as

errors in these measurements contribute first order

errors to the 2e\h measurement. Table 2 summarizes

the sources of uncertainty in 2e/h associated with the CIC. Since this instrument was not modified the uncertainties are essentially the same as reported and

discussed by FDL [1], Because the SPC and CIC are

of similar construction some of the sources of uncertainty for the CIC are the same as those for the SPC.

c) Null Detector Systems

Two independent null detector systems are used. Both systems consist of photocell galvanometer

amplifiers modified to drive a stripchart recorder. The

galvanometer amplifiers were: (a) modified to reduce thermal emf variations, (b) shielded electrostatically,

and (c) placed on a anti-vibration platform. The

amplifier used for the standard cell balance is operated

in the series feedback mode and modified for high

input impedance to reduce the off-null currents. The

amplifier used for the junction balances is operated in the parallel feedback mode to reduce the Johnson

noise. In the future we plan to use a superconducting

galvanometer (SQUID) to make the junction balances.

d) Standard Cell Comparison System

To compare the standard cells in the 2ejh labo- ratory with other cells, the cells are connected in

series opposition (negative leads connected together)

and the small voltage difference measured with a commercial potentiometer. The null detector used

with the potentiometer consists of a series feedback

photocell galvanometer amplifier driving a secondary

galvanometer.

The potentiometer system is regularly calibrated and a correction applied to the data. The systematic

uncertainty of the correction is estimated to be less than 10 ppm of the voltage difference being measured.

When an unsaturated cell is compared to a saturated

cell, the difference is about 1000 u.V and the systematic

uncertainty in comparing the emf's of two cells is less than 0.01 ppm. The potentiometer is used with a reso- lution of 0.01 u.V and the standard deviation of a single

observation is about 0.01 u.V, however, the use of a redundant statistical design reduces the random uncertainty by at least a factor of two.

5. Experimental Procedure

Prior to a 2ejh measurement, junction devices are

tested and a suitable one is mounted in the waveguide

holder. The device is replaced only if the junctions

become defective, otherwise they are left in the

cryostat between runs and are kept cold at liquid

nitrogen temperatures. On the day of the run the photocell amplifiers are connected to 12 V batteries and

allowed to stabilize. Standard cell comparisons are made using the commercial potentiometer, a photocell amplifier, and secondary galvanometer. The com- 220
parisons of cells are done using a redundant design of the type used in the NBS Volt Transfer Program [18]. Liquid helium is transferred and it is pumped below

2 K. The electronic equipment is turned on, micro- waves and bias current adjusted, and the system is left to stabilize for approximately 1.5 h. Afterwards

the bias current and frequency are adjusted to pre- viously selected optimum values and the thermal emf

in the measuring leads is checked. The 2ejh working group of cells contains three unsaturated standard cells mounted in a temperature

regulated enclosure. On any day's run one of these

cells is used throughout the entire run. The 1 V output

of the voltage comparator is balanced against the cell and the difference recorded on the strip-chart recorder. The low voltage output of the comparator is balanced

against the junction and recorded. During all balances the input to the null detector is reversed. The micro- wave frequency is measured immediately before and

after the voltage balance. The counter is set so that it does not sample during a junction balance as this

affects our measurements by disturbing the junctions.

Another balance of the cell is then made against the comparator. These balances are interpolated to the time of the junction balance to eliminate the effects

^ 483593 40

oi CM

1 1 1 21 JUN 72

DAY MEAN =483593 432GHj/VNBS -

G"= 0.031 ppm

<TU

- 0 0i4ppm

0 Ippm

• •

•

1 1 1300 1400 1500 1600 1700 TIME (HOURS)

Fig. 6. 2ejh results of a typical day's run. Each point consists of four balances against the device voltage to compensate for thermal emfs and series-parallel resistor string inequalities

of the linear drift of the current in the comparator.

The polarity of the bias current, the standard cell, and the comparator current is reversed and the pro- cedure is repeated. The series-parallel networks are interchanged and the procedure is repeated for each

polarity. The pattern of polarity reversals (+,—,—,+)

tends to cancel out the effects of first order drift of the thermal emf's in the measuring leads if the balances

are evenly spaced in time. The four junction balances (and associated cell balances) combine to give one

independent measurement of the standard cell emf in terms of 2ejh (or of 2ejh in terms of the standard

cell voltage). Typically four or five independent measurements are made during one day's run. During

the run small adjustments to the microwave power, microwave frequency, and bias current are made as necessary. At the conclusion of the run additional

standard cell comparisons are made between the

2ejh working group and the check standards. The result of a day's run consists of a recording of comparator balances and frequencies. Each balance

is adjusted close to null and straight lines can then be

fit by eye to the recording traces. A calibration signal

is produced by causing a known change ( ~ 1 ppm) in the current in the instrument to determine the nor- malized deviation. The normalized deviation repre- sents the fractional difference between the 100:1 comparator ratio and the actual standard cell-junction

ratio. One set of four comparator balances (including

polarity and series-parallel reversals) combines to give

(2e/A.).(78) = /S«v where F.s is the standard cell voltage, f> is the measured

voltage ratio VsjVj, Vj being the junction voltage, n is the step number, and v is the microwave frequency.

Using this equation, the product (2e/h • ( Vs ) can be

calculated from the 2cjh run. If either 2ejh or Vs is assumed known then the other can be calculated. A

plot of the results of a typical day's run is shown in Fig. 6. In February 1972 six saturated cells in a temperature regulated enclosure (check standard 1) were

placed in the 2ejh lab as a local voltage standard to estimate the precision with which a standard can be maintained via 2e/h measurements. This enclosure,

which was designed by Cutkosky [19], maintains a

stable temperature with short-term fluctuations of approximately 20 microdegrees Celsius. The mean emf

of the six cells in the enclosure is shown in Fig. 7. (Here a constant value of 2ejh has been assumed.) The

cell emf's are changing due to the large temperature

shock encountered when installed in the box and nor- mal cell aging, however, the emf's are very predictable

and the standard deviation of a linear least-squares

fitted line to the data in Fig. 7 is less than 0.02 fiV.

6. NBS Voltage Maintenance

and Dissemination Prior to July 1972 For many decades the U.S. legal volt has been

maintained by a large group of standard cells [20]. Over the years groups of cells made at NBS were

assembled and called the National Reference Group

(NRG). From 1955 to 1969 the NRG consisted of 44

saturated cadmium sulfate (Weston) cells. The mean

emf of the 44 cells was assumed to remain constant

in time. In 1963, monitoring of a second group of 18 saturated cells designated the Sesondary Reference Group (SRG) constructed in 1958 was begun. After

the move of NBS from Washington to Gaithersburg

in 1966, both groups of cells were put in two stirred temperature-regulated oil baths at 28 °C. In these

baths, both groups are exposed to the moving oil, and electrical connections to the cells are made by

opening the bath lid and inserting copper stabbers into mercury amalgamated copper cups. The temperature

of each oil bath is read with a calibrated platinum

resistance thermometer and Mueller bridge. The bridge

and thermometer are calibrated monthly, the latter at the triple point of water. By December 1969 the emf's of several cells in the NRG showed large instabilities due to the formation of gas bubbles. In order to maintain the stability of the

legal volt, a new mean for the NRG was calculated

221
(neglecting the unstable cells) using well established procedure [20], Measurements of the reference cells were made

using the "pivot cell" technique whereby all cells are read against only one cell (the pivot or favorite cell). In May 1970 a different procedure for comparing the

reference cells was started. This procedure provides

equal precision in determining any cell emf with

respect to the mean, as opposed to the pivot cell method which provides high precision in determining

the pivot cell and low precision in determining the

other cells with respect to the mean. At the same

time, a more accurate instrument with better resolu- tion (about 0.01 f!.V) was used for all comparisons.

Both methods of intercomparison were carried out

for several months until final conversion to the new

method was completed in September 1970. The new

method allowed greater flexibility in using any cells of the NRG or SRG for calibration purposes. In

practice, the Volt Transfer Program calibrations

prior to July 1972 were all made using a subset of 6 cells of the SRG. Since June 1970, all calibrations have been related to the NRG via the SRG subset.

the national units consisted of groups of saturated

cells, the international comparisons only gave in- formation on the relative drift of the various groups

of cells and not on the absolute stability. Until very

recently, the precision of the yp experiments permitted

checks of the stability of the NBS volt to about 2 ppm.

A possible annual drift in the mean emf of the NBS

reference cells as large as several tenths of a ppm could have gone undetected. Since 1970, sub-ppm deter- minations of 2ejh at the University of Pennsylvania

and NBS have provided checks on the stability of the NBS volt.

7. Determinations of 2e/h at NBS (1971—1972)

All determinations of 2e/h at NBS have been ob- tained by relating the Josephson junction voltage to the mean emf of the National Reference Group of standard cells. As discussed in Section V, direct 2e/h measurements were made on one of the cells of our working group. The emf of the working cell was re- lated to the mean emf of a second group of standard

cells by direct standard cell comparisons before and

after each 2e/h run. This second group of cells in turn

Fig. 7. The mean emf of six saturated cells assigned assuming

an arbitrary value of 2e/h. The cells are mounted in a temperature regulated enclosure located in the 2e/h laboratory

2 f/h IN TEAMS OF THE U S. LEGAL VOLT

~i—i—i—i—i—i—i—i—i—i—

Adopted Vo!u0 483593 <20 GHi/y

Jo,

_J I ' I I 1 I ' ' I I I I L

J J ASONDJFMAMJ J 1971 1972

Fig. 8. Values of 2ejh measured at NBS referenced to Vxbs, the legal volt. Each point represents 1 day's run which usually

consists of 3 to 5 independent determinations of 2e/h. The

open circle indicates a run in which the standard cell used for the measurements changed an excessive amount ( >0.1 u.V) during the run. [Error bars indicating within-run standard

deviation ( ~0.04 ppm) have been omitted for clarity.]

Two major improvements were made to one of the

oil baths : (a) installation of a new temperature regulating system, and (b) installation of a selector switch

for the SRG cells so that the bath lid would not have

to be disturbed. These two changes, made in March

1971 by Eicke, lowered the random uncertainty

of a standard cell comparison involving these groups by a factor of four [21]. Over the past 60 years, various methods have been

employed to place limits on the stability and reproducibility of the NBS volt. The two most meaningful

methods used prior to 1970 for determining these

limits were (a) the voltage comparisons of national

units at the Bureau International des Poids et Mesures (BIPM) conducted every 3 years [22]; and (b) nu- merous measurements of the gyromagnetic ratio of

the proton, yp , at NBS since 1958 [23]. Inasmuch as

was compared to the Secondary Reference Group

(which was directly assigned in terms of the NRG).

The second group of cells was necessary, since substantial scatter (~0.2 ppm) in the mean emf of the SRG prevented a precise check of possible changes in the working cell during a 2ejh run. A series of 32 2ejh measurements were made at NBS between July, 1971 and July, 1972 to monitor

the legal volt (i.e., the mean emf of the NRG) and to obtain sufficient data to permit a redefinition of the

U.S. legal volt directly in terms of the ac Josephson

effect. These 2ejh results are shown in Fig. 8. The

within-run standard deviation of the mean of a

typical point is about 0.025 ppm. A least-squares

linear fit to the data implies an apparent drift of about —0.41 + 0.03 ppm year in the legal volt during this period. Comparing the results of the last

222
determination of 2e/h at the University of Pennsylvania (May, 1970) with the first reported value at NBS (July to August, 1971), we find an apparent

drift of —0.22 + 0.14 ppm/year in the legal volt for 1970 to 1971 assuming the legal volt alone has changed

during this period. The relative agreement of these two numbers suggests that the U.S. legal volt has been

decreasing for the period 1970 to 1972. For the NBS measurements (see Fig. 8), various changes were made from time to time in the proce- dures used to relate the Josephson volt to the legal

volt. These changes primarily involved the use of

different standard cells. For runs 1 through 14, the

2e/h working cell was one of three saturated cells mounted in a commercial temperature-regulated

enclosure at 30 °C. For subsequent runs, the working

cell was one of three unsaturated cells mounted in the same enclosure. The unsaturated cells were used

because changes in emf of the saturated cells, due

ostensibly to temperature changes of the enclosure, were limiting the within-run precision of the 2e/h measurements. Standard cell comparisons made

before and after the 2e/h runs indicated a typical

change of about 0.04 u.V for the saturated cells but

a change of only 0.02 u-V for the unsaturated cells. Several different intermediate groups were used

to relate the 2e/h working group to the Secondary

Reference Group of cells. All intermediate groups, the SRG, and the NRG were located in a separate shielded room next to the shielded room in which the 2ejh measurements were carried out. For the early 2ejh runs (prior to September, 1971), an air enclosure con- taining four saturated cells was used as the inter- mediate group. This intermediate group was compared to the SRG weekly but not necessarily on the same days as 2e/h runs. Since September 1971, an

NBS-built air enclosure containing six saturated cells (which we define as "check standard 2") has been

used as the intermediate group. In Fig. 9, we have

assumed an arbitrary value of 2ejh and plotted the

effective mean emf of four of the cells using our 2e/h data. The standard deviation of a single point about

the fitted line is 0.04 ppm. After January, 1972,

comparisons of this intermediate group with the SRG

were made on the same day (while the 2ejh run was in progress) rather than on arbitrary days. In order to facilitate the comparisons of large

numbers of standard cells, we installed several rotary

selector switches which permitted rapid connection of

different cells (from the various groups) to the potentio- meter. These switches, which were operated in air, were electrostatically and thermally shielded. The

thermal emf's present in these switches were on the order of a few nanovolts. The sources of uncertainty in our 2e\h determinations can be divided into two categories: (a) the random and systematic uncertainties associated with

the 2ejh measurements, and (b) the uncertainties in relating the voltage of the 2ejh working cell to the

legal volt (i.e., the NRG). A summary of these sources of uncertainty and the corresponding one-standard- deviation estimates is listed in Table 3. The following

comments apply. The random uncertainty arises from

random variations in the thermoelectric voltages,

day-to-day fluctuations in the local and reference

CHECK STANDARD 2 [le/h - 48359 3 J 20 Cm

-i 1 T

J 1 I I I I I L

S O N D J F M

1971

Fig. 9. The mean emf of four cells, assigned assuming an

arbitrary value of 2ejh. This enclosure is located in a separate

shielded room containing the NBS reference group of cells

(i.e. the intermediate, SRG, and NRG) groups of cells and any other randomly varying experimental parameters present in a 2e/h experiment. The standard

deviation for a single run referenced to the legal volt

(i.e. NRG) was of order 0.04 ppm. The estimated uncertainty in the measurement of the frequency of the

applied radiation is 5 parts in 109 and thus essentially

negligible. The systematic uncertainty in relating the

Table 3. Summary of sources of uncertainty in 2e/h for April 1972 Uncertainty parts

in 108 (one standard

deviation estimates)

1. Measurement uncertainties

a) Random uncertainty of the 0.9 predicted value

b) Frequency measurement and 0.5

stability

c) Low-voltage comparison 2.8

(series-parallel comparator)

2. Local volt uncertainty in Fnbs«» 3

(April 1972) RSS total 4

local volt (the 2e/h working cell) to the legal volt has been ascribed to thermoelectric emf's in the measuring

leads, temperature variations in the standard cell enclosures, and possible undetected changes in the working cell during 2ejh measurements. A value of 2ejh based on the first nine runs shown

in Fig. 8 has been reported [12]. The result was:

(2e/&),= 483593.598 ± 0.024 GHz/F^ (0.05 ppm) . The random uncertainty of the mean of these nine measurements was 2.3 parts in 109

. This determination was related to the results of other workers [2, 13, 14] via direct standard cell comparisons carried out

223
between NBS and other national laboratories under

the auspices of B1PM during the summer of 1971. In the spring of 1972, a second series of similar volt comparisons was carried out. A value of 2ejh referenced

to the legal volt for April, 1972 (approximately the

central date of this latter series of comparisons) has been obtained by linearly fitting all the 2ejh data from

January to July, 1972. The result is:

(2e//t)„= 483593.444 ± 0.019 GHz/7NBS (0.04 ppm) . The random uncertainty of 9 parts in 109 (as indicated

in Table 3) was obtained by calculating the standard

deviation of the interpolated result. The Josephson voltage is directly proportional to the frequency of the applied radiation with the pro- portionality constant precisely equal to the physical constant 2ejh. Since no other method of determining

this physical constant with an accuracy comparable to that of the ac Josephson effect exists, we can use it only to establish a very stable and precise voltage standard but not to establish an absolute one. In

order to implement such a standard at NBS, we have

decided to adopt a value of 2e/h consistent with the assumed value of the mean emf of the National Reference Group on July 1, 1972. This value of 2ejh was obtained by fitting the 2ejh data shown in Fig. 8

the relationships betweon the national units must be

determined by comparison experiments. Traditionally

these comparison experiments are conducted trien- nially at BIPM (Sevres, France). In order to better determine the relationships of the units in laboratories carrying out high precision 2ejh experiments, two

series of direct volt transfers between NBS and each

of these laboratories were completed about 8 months

apart. An individual transfer experiment involved the

following: (a) calibration of an air enclosure in teims

of the NBS reference cells and shipment of the temperature-controlled enclosure to the second laboratory,

(b) calibration of the enclosure at the second labora- tory and shipment back to NBS, and (c) calibration at NBS. Each transfer experiment lasted about

6 weeks. These transfer experiments provide a good

basis for comparing the values of 2e/h obtained at the various national laboratories. In Fig. 10, we have plotted values of 2e/h obtained

by the various workers applicable to the time2 of the

transfers [2, 13, 14]. The results presented have been

normalized to the NBS volt (^NBS ) using the data

from the direct volt comparisons. Each error bar

indicated the one standard deviation uncertainty

assigned by the respective workers in determining 2ejh

relative to their particular national unit of emf. No

SUMMER

1971 SPRING

1972

TIME ( MON T H S I

Fig. 10. Values of 2e/h as determined by workers at various

national laboratories. All values have been expressed in the NBS unit of voltage ( Fnbs) using the results of direct standard

cell comparisons between each laboratory and NBS. (See Ref. [24].)

from January to July, 1972. The (extrapolated) result

is:

(2e/fc)m = 483593.420 ± 0.019 GHz/FNB8 (0.04 ppm) . The uncertainties in this result are essentially indentical to those of the April value.

8. Comparison of Recent Values of 2e/h (1971—1972)

Workers at various national 1 aboratories have con- ducted and reported experimental determinations of

2e/h referenced to their particular national unit of volt- age. Each of these units has been denned as the mean

emf of a different group of standard cells and therefore

National

Referc Group

C e 6 3»

2e//i wor k ing group

check

standard

1

check

standard

2

VTP

working

group

cali bra tion wor king group

VT P enclosures

regular

calibration work load

Fig. 11. A block diagram outlining the principle steps in the

present (1972) NBS voltage maintenance and dissemination program using 2e/h via the ac Josephson effect

additional allowance for the uncertainty of the volt

transfers has been added to the workers' estimated

2 The PTB result shown for the summer of 1971 is based on a 2e/h determination made about l /a year earlier and is the only reported value available for comparison with the 1971

direct standard cell transfer experiment.

224
uncertainties. The uncertainty in each of these volt

transfer experiments has been estimated to be about

0.2 ppm [24]. The results are in fair to good agreement

when the additional uncertainty associated with each

volt transfer is included. Critical comparisons of pre- sent (1972) determinations are limited by the transfer of standard cell enclosures. The development of an improved portable voltage standard (for example, a

portable Josephson standard) should permit meaningful comparisons of 2e/h experiments at different laboratories with transfer uncertainties comparable

to local 2elh uncertainties of a few parts in 108

.

9. NBS Volt Maintenance Via the Josephson Effect In July, 1972, NBS began maintaining and disse- minating a unit of emf based on the ac Josephson

effect. To define this new unit it was necessary to choose a value of 2ejh. This new unit was defined to be

equal to the apparent mean emf (on July 1, 1972) of

the National Reference Group of standard cells. The

value of 2ejh chosen for this purpose is the NBS 2ejh

result for July, 1972 previously discussed in section VIII, namely

2e/A= 483593.420 GHz/ .

Thus, the NBS as-maintained unit of voltage is now

directly related by this constant (which is assumed to be exact) to the unit of frequency.

The procedures used to maintain and disseminate

the new unit are essentially the same as those used in the 2e/h measurements already described in this paper. A diagram indicating the principle steps in the NBS

2e/h voltage dissemination program is shown in Fig. 11. Periodically, the Josephson apparatus is used to "calibrate" the emf of one of the standard cells in the

2e/h working group. This cell is compared with two

other groups of cells (check standards 1 and 2 in

Fig. 11) immediately before and after the 2e\h measurements. If the observed within-run standard deviation

of the 2e/h measurement is less than twice the ex- pected value 2a (presently a ^ 0.04 ppm), the assigned

emf of the 2e/h working cell is used to obtain the emf's of the cells in the two check standards. The mean emf of each check standard is plotted versus time (as shown in Fig. 7 and 9). The 2e/h measurements are made at regular intervals (about every 2 weeks for the data presented here) and mean

emf's of check standard 2 and a second (VTP) working

group of cells are obtained for use in calibrating the

various transport standards and other groups of cells by averaging the respective results of the five most

recent 2ejh runs. For calibration purposes, values for these mean emf's are calculated for up to several weeks ahead by applying small corrections for the

long term (3 to 6 months) drift of the check standard and working group means3

.

3 Generally, an average of five 2e/h runs is used; however,

this number may change depending on the time intervals be- tween 2e/h runs. The data from a given 2e/h run are only used

for calibration if the corresponding value for each check stand- ard mean is within 0.1 ppm of the expected value for the mean, and analysis of the individual cell emf's of the check standards shows no anomalous behavior. The small drift corrections (which may be negligible) are applied to the check standard and working group averages (for calibration purposes only) on

a weekly basis.

10. Conclusions The precision with which the mean emf of a single group of standard cells can be maintained using the present Jospehson apparatus is about 2 parts in 108 and is a measure of how reproducibly we can calibrate a group of standard cells over an extended period of

time. The accuracy with which the emf's of a group of

cells can be determined in terms of a unit based on an

adopted value of 2e/A is about 4 parts in 108 and

includes allowances for systematic effects in the pre- sent measurement system. This is an estimate of how

well the present system would agree with another

totally independent 2ejh system.

The agreement between the NBS 2ejh system and

systems in operation at other national laboratories has been tested indirectly via the shipment of stand- ard cells. The results are inconclusive in determining

the relative agreement of the various 2ejh measuring

systems at their current level of precision ( < 0.1 ppm).

Differences as large as 0.4 ppm between 2e/h results and the corresponding cell transfers have been observed. These differences have been attributed pri- marily to uncertainties in the standard cell transfers. One way to compare 2e/h systems at precisions

approaching a few parts in 108 is to use a portable 2e/h system rather than standard cells. Our present system

(particularly the dc instrumentation) is portable and

can be used in such a comparison experiment. The

application of cryogenic techniques to dc instrumentation (e.g. superconducting galvanometers and cryo- genic dividers) and the development of an improved

(long lived and durable) Josephson junction should permit the development of a compact cryogenic 2ejh

system. Finally, with the use of cryogenic instrumentation, it appears that the accuracy of the 2e/h voltage ratio measurements can be improved at least an order of magnitude.

Acknowledgements. We would like to thank T. J. Witt for his assistance in setting up the present 2ejh system and his collaboration in the initial series of NBS measurements, and

D. N. Langenberg and the University of Pennsylvania for the use of several instruments and a standard cell enclosure. We

would also like to thank members of the Absolute Electrical Measurements Section, particularly W. G. Eicke, H. H. Ellis, and B. N. Taylor, for assistance.

References

1. Finnegan, T. F., Denenstein, A., J*angenberg, D. N. : Phys. Rev. B 4, 1487 (1971).

2. Harvey, I. K., Macfarlane, J. C, Frenkel, R. B.: Metrologia 8, 114 (1972); and private communication.

3. Clarke, J.: Phys. Rev. Letters 21, 1566 (1968).

4. Clarke, J.: Phys. Rev. Letters 28, 1363 (1972).

5. Bracken, T. D., Hamilton, W. O.: Phys. Rev. B 6, 2603

(1972).

6. Bloch, F.: Phys. Rev. Letters 21, 1241 (1968); Phys. Rev. B 2, 109 (1970).

7. Rieger, T. R., Scalapino, D. J., Mercereau, J. E. : Phys. Rev.

Letters 27, 1787 (1971).

8. Tinkham, T., Clarke, J.: Phys. Rev. Letters 28, 1366

(1972); Tinkham, M.: Phys. Rev. B 6, 1747 (1972).

9. Fulton, T. A.: Phys. Rev. (to be published).

10. Parker, W. H., Langenberg, D. N., Denenstein, A., Taylor, B. N.: Phys. Rev. 177, 639 (1969).

11. Taylor, B. N., Parker, W. H., Langenberg, D. N., Denenstein, A.: Metrologia 3, 89 (1967). 12. Finnegan, T. F., Witt, T. J., Field, B. F., Toots, J:. In: Proo. 4th Int. Conf. on Atomic Masses and Fundamental

Constants, p. 403 (Sanders, J. H., Wapstra, A. H., Eds.). New York: Plenum 1972.

225
B. F. Field et al. : Volt Maintenance at NBS via 2e/A; A New Definition of the NBS Volt

13. Kose, V., Melchert, F., Faok, H., Hetzel, W.: IEEE Trans. Instrum. Meas. IM-21, 314 (1972); and private communication.

14. Gallop, J. C, Petley, B. W. : Metrologia 8, 129 (1972); and

private communication.

15. Stephen, M. J.: Phys. Rev. 81, 531 (1969).

16. Kose, V. E., Sullivan, D. B.: J. Appl. Phys. 41, 169 (1970).

17. Wenner, F.: J. Researoh Natl. Bur. Standards 25, 229

(1940).

18. Volt Transfer Program Instructions. Electricity Division, National Bureau of Standards, 1971 ; see also Eioke, W. G., Cameron, J. M.j U.S. Nat. Bur. Stand. Technical Note 430

(1967).

19. Cutkosky, R. D. (unpublished).

20. Hamer, VV. J.: U.S. Nat. Bur. Stand. Monographs 84

(1965).

21. Eioke, W. G.: Private communication.

22. Terrien, J. : Private communication. See also Metrologia 7, 78 (1971).

23. Driscoll, R. L., Olsen, P. T.: In: Proc. Int. Conf. on Precision Measurement and Fundamental Constants, p. 117 (Langenberg, D. N., Taylor, B. N., Eds.). U.S. Nat. Bur.

Stand. Spec Pub. 343, 1971.

24. Eicke, W. G., Taylor, B. N.: IEEE Trans. Instrum. Meas.

IM-21, 316 (1972).

226

USCOMM-NBS-DC
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-21, NO. 4, NOVEMBER 1972 Summary of International Comparisons of

As-Maintained Units of Voltage and

Values of 2e/h

WOODWARD G. EICKE, JR., and BARRY N. TAYLOR

Abstract -Using temperature-regulated transportable standard-cell enclosures, the National Bureau of Standards (NBS), under the auspices of the Bureau International des Poids et Mesures (BIPM), during the period June 1971 through June 1972 has carried out a series of direct comparisons of the units of voltage as-maintained by NBS and BIPM,

Sevres, France; the National Physical Laboratory, United Kingdom; the National Research Council, Canada; the National Standards Laboratory,

Australia; and the Physikalisch-Technische Bundesanstalt, Germany. The main purpose of these comparisons was to provide a sound basis

for intercomparing values of 2e/h obtained at the various national

laboratories via the ac Josephson effect in superconductors. It was

found that when converted to a common unit of voltage, most measured

values of 2e/h agreed with each other to within the 1 to 2 parts in 107 estimated uncertainty (1 standard deviation) of the volt comparisons. This satisfying result would seem to indicate that serious consideration should be given to adopting a single international value of

2e/h for use in maintaining units of voltage.

I. Introduction

THE UNIT of voltage is generally maintained in the various national laboratories via large reference groups of saturated standard cells kept at constant temperature and for which the mean emf of the group is assumed constant with time. In order

to maintain worldwide consistency in the various as-maintained

units of voltage and to provide information on their relative

drift rates, the Bureau International des Poids et Mesures (BIPM)

conducts, at about three year intervals, international comparisons of the units maintained by the national laboratories. These comparisons are carried out in the following way [1]. Each participating national laboratory sends to BIPM a group

of saturated standard cells which have been calibrated in terms of its own as-maintained unit. The cells are sent in nontemperature-regulated containers and are often hand carried. At BIPM, the cells are placed in a common oil bath with the BIPM reference group of cells, allowed to stabilize for several months, and then intercompared among themselves. Finally,

they are returned to the originating national laboratories where

they are recalibrated in order to determine how much the cells have changed during the overall period, which could be as long

as six months or more. Often, changes in individual-cell EMF's

as large as 1 or 2 /iV are observed; nevertheless, the mean of the

cell assignments before and after shipment is taken as the mean EMF of the cells while at BIPM.

Manuscript received June 24, 1972: revised July 14, 1972. The authors are with the Absolute Electrical Measurements Section,

Electricity Division, Institute for Basic Standards, National Bureau of

Standards, Gaithersburg, Md.

In early 1967, in cooperation with the U.S. Air Force, the National Bureau of Standards (NBS) undertook a program to improve the interlaboratory transfer of the unit of voltage within the United States. As a result of this program, it was

found that the transfer process could be substantially im- proved by the following.

1) Having NBS ship to the client in question an NBS-owned

commercially available standard-cell enclosure with shippable

saturated standard cells and maintained under continuous

temperature control. (Power is supplied by batteries while in transit.) The cells are calibrated in terms of the NBS unit

prior to shipment, compared by the client to the cells he uses

to maintain his unit (with the same equipment and methods he

uses in his everyday work), and returned to NBS for recalibration, still under temperature control. 2) Carefully controlling the transportation of the enclosure between NBS and the laboratory in question. This may be accomplished via a special shipping container and proper ship- ping arrangements. (Most enclosures are shipped via air freight.) 3) Using statistically sound experimental designs [2] and

procedures for making the intercomparison measurements and

for analyzing the resulting data. The standard deviation for a

single transfer using these techniques was found to be between

1 and 2 parts in 107. As a result of the success of the initial program, a new cali- bration service based on the above methods was developed and termed the volt transfer program (VTP). [It is now part of the NBS measurements assurance program (MAP).]

During the development of the VTP within the United

States, the question of its use for international comparisons

naturally arose. In 1968, NBS explored this possibility with BIPM. Subsequent transfers between NBS and BIPM showed

that the vagaries of international flights and zealous customs

inspectors not withstanding, VTP techniques could be successfully used to transfer the unit of voltage internationally.

Indeed, NBS used the results of such a transfer for the 1970

triennial international comparison at BIPM [1]

.

During the time period in which the VTP was being de- veloped, many national laboratories began work on the ac Josephson effect in superconductors with the idea of eventually

using measurements of 2e/h to maintain their unit of voltage. This work has now progressed to the point where 2e/h via the

ac Josephson effect can be determined in terms of a particular as-maintained unit of voltage to 1 part in 10 7 or better. This

level of accuracy significantly exceeds the accuracy of the differences in the units of voltage maintained by the different

227
EICKE, JR., AND TAYLOR: COMPARISONS OF VOLTAGE UNITS AND VALUES OF 2e/h

TABLE I Summary of 1971 and 1972 International VTP Transfers

Approximate EMF's at NBS Assumed Mean Before After EMF at EMF at Enclosure Date Transit Transit Lab.,£N BS Lab..£NBS ^LAB " KNBS

Lab. Number at Lab. OjKnbs) (jjKnbs) (mKnbs) (»Vlab) (mKnbs)

BIPM 1100 30 Nov 71 138.85 138.95 138.90 139.19 -0.28 BIPM 1300 15 May 72 142.91 142.65 142.78 143.01 -0.22 NPL 1100 31 Jul 71 138.61 138.45 138.53 137.38 +1.13 a NPL 1300 28 Mar 72 142.74 142.91 142.82 141.75 +1.07 NRC 1400 1 Mar 72 143.35 143.08 143.21 143.72 -0.49 NSL 1100 18 Jun 71 138.44 138.61 138.54 138.07 +0.45 a NSL 1500 3 Apr 72 155.66 155.70 155.68 155.29 +0.38b PTB 2200 24 Aug 71 34.52 34.43 +0.09c PTB 1500 15 May 72 156.22 156.27 156.24 156.29 -0.05 3

Notes: 1) All EMF's reduced by 1 018 000 fiV and corrected to the nominal enclosure temperature.

2) The nominal temperature for all enclosures was 30°C except for 2200 which operated at 32°C. ?One cell excluded. Two cells excluded.

c Simple means not applicable due to nonlinear drift.

national laboratories as obtained from the triennial inter- national comparisons at BIPM. Thus, there is no reliable way

of assessing the agreement or lack thereof between the several different measurements of 2ejh, and consequently the potential of the Josephson effect for maintaining units of voltage. To alleviate this problem, that is, to provide a sound basis

for comparing values of 2elh obtained at the various national

laboratories, and also to provide further experience to BIPM

and other laboratories in using VTP techniques, NBS initiated

in 1971, under the auspices of BIPM, an experimental program

to apply VTP methods to the international comparison of

as-maintained units of voltage. During the period June 1971

through June 1972, transfers were carried out between NBS

and BIPM, Sevres, France; the National Physical Laboratory

(NPL), United Kingdom; the National Standards Laboratory

(NSL), Australia; the National Research Council (NRC),

Canada; and the Physikalisch-Technische Bundesanstalt (PTB),

Germany. All of these laboratories participate in the triennial international comparisons and have (or will shortly have)

active Josephson 2e/h volt-standard programs. Where appli- cable, the researchers carrying out the Josephson 2e/h measure- ments were encouraged to make 2e\h determinations during

the period the NBS transport standard was at their laboratory

in order to tie together all of the 2e/h measurements as closely as possible.

II. Experimental results

Two distinct comparisons were carried out, the first pri- marily during the summer of 1971, and the second during the

spring of 1972. The procedures used were essentially those described in Section I for the NBS VTP except that each

national laboratory calibrated the transport standard using its own normal procedures. The transport standards, each containing four cells, were shipped via air freight to and from

each laboratory, with the standard remaining at the laboratory

from two to four weeks. When a serious problem was en- countered with a particular transfer, it was repeated as quickly

as possible. The difference between the units was calculated using the relationship

LAB

» r _ {E )lab ~ {-^Inbs Tr ~ 'NBS ~ ~ rri 'NBS {E} (1) LAB

where {iT} lab and {^Inbs are tne numerical values only of

the EMF assigned at the laboratory and NBS, respectively, and FLAB and VNBS are the respective units of voltage. The

results of the two comparisons are summarized in Table I. The assumed standard deviation uncertainty for a single transfer is 0.14 juV, that adopted for the VTP in the United States. On three occasions, erratic cell behavior required doubling the

variance (i.e., the assumed uncertainty for PTB 71 and 72 and NSS 72 was taken to be 0.20 )uV). Table II summarizes the relevant 2e/h measurements carried out during the period of the direct volt transfers. Two of the

participating laboratories, BIPM and NRC, were just getting

their 2e/h work underway and did not have any results to re- port. NBS [3] ; NPL [4] ,[5] ; NSL [6] ,[7] ; and PTB [8]

all had 2e/h programs in operation and obtained a value of

2e/h in terms of their as-maintained unit during the time the transport standard was in their laboratory. (Unfortunately, a 2e/h measurement at PTB could not be completed during the summer of 1971 and the fall 1970 result must be used.) All uncertainties are meant to correspond to 1 standard deviation and are those given by the experimenters involved; generally,

they contain both random and systematic components of

uncertainty.

The last column in the table gives the change in Flab from

1971 to 1972 A VLAB as implied by the 2e/h measurements. In principle, the uncertainty to be assigned a particular A FlAB should simply be the root-sum-square of the random uncertainty of the 1971 and 1972 measurements,

since the systematic uncertainty is common to both. The

reliability of the values for AFLab should, therefore, be

significantly better than is implied by the total uncertainty

assigned the individual 2ejh values. With this in mind, it may

be concluded that significant changes in some of the as-maintained national units of voltage have taken place. Table III summarizes the differences in the units of voltage maintained by the participating national laboratories and NBS

228
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, NOVEMBER 1972

TABLE II Summary of 2e/h Measurements via AC Josephson Effect

1971 1972

Implied

Change in ^LAB from

1971 to 1972, AKlaB

Lab. (MKLAB) 2e/h (GHz/KLAB)

Uncertainty

(ppm)

Approximate Time

Period of Measurements2ejh

(GHz/VLAB)

Uncertainty

(ppm)

Approximate Time

Period of Measurements

NBS NPL

NSL

PTB

483 593.589(24)* 483 594.15(10)c 483 593.80(5)e 483 593.7(2)80.05

0.2

0.1 0.4

July-Aug. 1971

July 1971

June-July 1971

Fall 1970

483 593.444(24)b 483 594.00(10)d 483 593.733(48)f 483 593.606(20) h

0.05

0.2

0.1 0.04

Apr. 1972

Apr. 1972

Mar.-Apr. 1972 May 1972

-0.30

-0.31 -0.14 -0.19

Note: Numbers in parentheses are estimated 1 standard deviation uncertainties in last digit(s) of main number.

"See [3]. ^See[5]. *See[7]. *See[8].

b See[10]. dSee[ll]. fSee[12]. hSee[9].

TABLE III Summary of Differences in Units of Voltage Maintained by Participating National Laboratories and NBS

International Direct Volt Via Values of Direct Volt Via Values of Comparison8 Transfers 2e/h Transfers 2efh ^lab- KNBS (m^bipm) O^NBS) (m^nbs) O^NBS) (M^NBS)

^BIPM ' - VNBS -0.17 -0.28 ± 0.14 -0.22 ± 0.14 VNRC ~ KNBS -0.07 -0.49 ± 0.14 ^npl - fNBS +0.52 +1.13 ± 0.14 + 1.16 ± 0.21 +1.07 ± 0.14 + 1.15 ± 0.21 ^nsl- kNBS -0.17 +0.45 ±0.14 +0.44 ± 0.11 +0.38 ± 0.20 +0.60 ±0.11 KPTB - KNBS -0.43 +0.09 ± 0.20 +0.23 ± 0.40 -0.05 ± 0.20 +0.33 ± 0.06

aCentral date, Feb. 1, 1970; see [1] . Note that KBIPM and ^NBS dil

"fer by less tnan !/ lu6 and

therefore small volt differences are for all practical purposes the same whether expressed in terms

of ^BIPM or KNBSas determined via the 1970 triennial international comparisons

at BLPM, the direct volt transfers carried out in 1971 and 1972

(Table I), and the 2e/h measurements as given in Table II. The 2e/h volt differences were calculated from the following

readily derived equation which is analogous to (1):

v v -

{ 2c } lab ~ {2e/ft} NBS t . ,~ 'LAB V NBS ~ r~ ,,\ y NBS \£)

where { 2e/h } is the numerical value only. 1

III. Conclusions

At least two important conclusions may be drawn from the data of Table III.

1) The 1971 direct volt transfer differences and those implied by the 2e/h measurements are in extraordinarily good

agreement (compare columns 3 and 4). In all three cases, the disagreement is significantly less than what would be expected from the assigned uncertainties. For the 1972 transfers, the

results are quite consistent with the assigned uncertainties (compare columns 5 and 6), but the agreement is not quite as satisfactory as for 1971. In view of the fact that the transport standards were not as well behaved during the 1972 transfers,

this result is not too surprising.

'Table III is equivalent to converting all of the 2e/h measurements

to a common unit of voltage and intercomparing the resulting values. We prefer the present method because the numbers are easier to digest.

2) There is an apparent difficulty with the value for ^bipm " ^nbs as obtained via the 1970 BIPM triennial comparison. This becomes evident when, for example, the triennial differences are compared with either the 1971 direct transfer differences or 2e/h differences (compare column 2 with either columns 3 or 4). With the exception of the 1971 direct BIPM

transfer, there are strong indications that the 1970 triennial

result for FBi PM - FNBS is too small by about 0.6 juV. On the other hand, it is difficult to understand why the 1971 (and also 1972) direct transfers between BIPM and NBS do not confirm

this apparent discrepancy but rather support the 1970 result. We leave further analysis of the data of Table III for a future

publication.

In summary, we may conclude the following from the work

reported here.

1) VTP type transfers can be successfully used internationally. The as-maintained units of voltage of different countries may be quickly and reliably compared using shippable standard cell enclosures maintained under continual temperature control. In our opinion, this method of comparison is a significant improvement over the traditional one and

should be the technique adopted for use in all future BEPM

triennial international comparisons.

2) Measurements of 2e/h at the various national laboratories

are well in hand.

3) In view of the agreement among the 2e/h measurements,

serious consideration should be given to adopting a single

229
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 1M-21, NO. 4, NOVEMBER 1972

international value of 2e/h for use by all those laboratories

that wish to maintain their units of voltage via the Josephson

effect. 4) There is a definite need for developing improved voltage reference and transport standards in order to take full advantage of the inherent accuracy of the 2e/h measurements. In our opinion, difficulties alluded to above are due to the in- adequacies of these voltage standards. Determinations of

2e/h are clearly pushing the existing technology in this area. Acknowledgment

The results reported here were made possible only by the dedicated efforts of a large number of individuals at the parti- cipating laboratories. At the risk of overlooking someone, we

should like to gratefully acknowledge the collaboration of the following persons: J. Terrien, Director, BIPM, under whose

auspices the transfers were carried out; G. Leclerc, BIPM;

J. J. Denton, J. C. Gallop, and B. W. Petley, NPL; A. F. Dunn

and G. H. Wood, NRC; R. C. Richardson, I. K. Harvey, J. C. Macfarlane, and R. B. Frenkel, NSL; W. Hetzel, V. Kose,

F.Melchert, andH. Fack, PTB; and B. F. Field, T. F. Finnegan,

J. Toots, and T. J. Witt, NBS. We are particularly grateful to H. H. Ellis, NBS, for making the hundreds of required measure- ments on the transport standards. We should also like to thank

Miss Thelma Gable of Pan American Airways for her assistance

in arranging for the shipment of the standards.

References

[1] G. Leclerc, "Rapport sur la I2eme comparaison des etalons na- tionaux de force electromotrice," Bur. Int. Poids Mes., Sevres, France, 1970.

[2) W. G. Eicke, Jr., and J. M. Cameron, "Designs for the surveillance of a small group of standard cells," NBS Tech. Note 430, 1967.

[3] T. F. Finnegan, T. J. Witt, B. F. Field, and J. Toots, "Measure- ments of 2e/h via the ac Josephson effect," in Proc. 4th Int. Conf. Atomic Masses and Fundamental Constants, J. H. Sanders and A. H. Wapstra, Eds., to be published.

[4] J. C. Gallop and B. W. Petley, "Recent NPL work on 2e/h," in Proc. 4th Int. Conf. Atomic Masses and Fundamental Constants,

J. H. Sanders and A. H. Wapstra, Eds., to be published.

[5] B. W. Petley and J. C. Gallop, "Measurement of 2elh by the Josephson effect," in Precision Measurements and Fundamental

Constants, D. N. Langenberg and B. N. Taylor, Eds., NBS Spec.

Publ. 343, U.S. GPO, Washington, D.C., pp. 227-229, 1971.

[6] I. K. Harvey, J. C. Macfarlane, and R. B. Frenkel, "Determination

of lelh based on the ac Josephson effect," Phys. Rev. Lett., vol. 25, pp. 853-856, 28 Sept. 1970.

[7] —, Monitoring the NSL standard of emf using the ac Josephson

effect," Metrologia, to be published.

(8] V. Kose, F. Melchert, H. Fack, and H.-J. Schrader, "Die Bestim- mung von e/h mit hilfe des Josephson-effektes," PTB Mitt.,

vol. 81, pp. 8-10, Jan. 1971.

(9] V. Kose, PTB, Germany, private communication.

[10] B. F. Field, NBS, Washington, D.C., private communication.

[11] B. W. Petley, NPL, United Kingdom, private communication.

[12] I. K. Harvey, NSL, Australia, private communication.

Reprinted by permission from IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT

Vol. IM-21, No. 4, November 1972, pp. 316-319 Copyright 1972, by the Institute of Electrical and Electronics Engineers, Inc. PRINTED IN THE U.S.A.

230
A FLEXIBLE SYSTEM WITH TWO SELECTABLE RATIOS FOR USE WITH JOSEPHSON DEVICES,

David W. Braudaway, IEEE Trans. Instrura. Meas. IM-27, December 1978.

Described is a flexible ratio and detection system for comparing dc

voltages developed by means of the ac Josephson effect at either 5 or 10 mV

to standard cell voltage. The system is self-calibrating in establishment of

the two ratios. A fan-wit hin-a-fan arrangement for current and potential

leads is employed to enhance the low value series-parallel resistor combinations. Normal operation is in a deflection mode in which the difference

between standard cell voltage and ratio set voltage is read on a high- impedance digital microvoltmeter . Based on root-sum-square combination,

the system uncertainty is 5 in 10®. Also described are the results of an intercomparison between the Sandia 2e/h volt, and the NBS 2e/h volt which

indicates no significant difference. Employed in the intercomparison was a specially modified standard-cell air-bath which was shipped by air.

HIGH ACCURACY POTENTIOMETERS FOR USE WITH TEN MILLIVOLT JOSEPHSON DEVICES.

II. CASCADED-INTERCHANGE COMPARATOR, A. Denenstein and T. F. Finnegan

(Department of Physics and Laboratory for Research on the Structure of

Matter, University of Pennsylvania, Philadelphia, PA 1917*0, Rev. Sci.

Instrum. 45, 735-741 (1974).

A potentiometer technique for accurately determining a fixed voltage

ratio has been developed which employs a voltbox subdivided for optimum

self-calibration. The calibration procedure minimizes the effects of resistor self-heating and interconnection resistances which are commonly limiting

factors. The basic theory of the technique and the design and construction

of an instrument to compare a Josephson device (10 mV) and a standard cell

(1 V) are described. This cascaded-interchange comparator (CIC) was evaluated in conjunction with another instrument, based on the double

series-parallel exchange technique, in a series of 2 e/h measurements and

direct instrument comparisons. The over-all uncertainty for the CIC was

about 3 parts in 10® (1o) and the direct comparisons between the two instruments agreed to within their combined uncertainty of 4 parts in 10®. The

cascaded-interchange technique should also be useful for the accurate measurement of high voltages.

231
HIGH ACCURACY POTENTIOMETERS FOR USE WITH TEN MILLIVOLT JOSEPHSON DEVICES.

I. DOUBLE SERIES—PARALLEL EXCHANGE COMPARATOR, T. F. Finnegan and

A. Denenstein (Department of Physics and Laboratory for Research on the

Structure of Matter, University of Pennsylvania, Philadelphia, PA 19174),

Rev. Sci. Instrum. 44_, 944-953 (1973).

High accuracy determinations of 2e/h using the ac Josephson effect

require correspondingly accurate measurements of small dc voltages. In the

preliminary stages of a determination of 2e/h with substantially improved

accuracy, two different potentiometric instruments based on two independent

principles of operation were designed and built. Both instruments were used

in an extended series of 2e/h measurements at the University of Pennsylvania

and subsequently at the National Bureau of Standards where they have been

used to realize the U.S. legal volt as recently redefined via the ac Josephson effect. One of these instruments is based on a modified series-parallel

technique and is described in detail in this paper. After treating the

general theory of the low resistance case, the design and construction of a portable self-checking instrument based on the double series-parallel

exchange method is described. This series-parallel comparator (SPC), built

to compare a Josephson device (10 mV) and a standard cell (1 V), was evaluated and tested in the course of determinations of 2e/h with one standard

deviation (1o) uncertainties of about 0.1 ppm. The sources of systematic

uncertainty are discussed. The uncertainty attributable to the comparator

alone is about 3 parts in 10^ (1o).

232
ac-JOSEPHSON-EFFECT DETERMINATION OF e/h: A STANDARD OF ELECTROCHEMICAL

POTENTIAL BASED ON MACROSCOPIC QUANTUM PHASE COHERENCE IN SUPERCONDUCTORS,

T. F. Finnegan, A. Denenstein, and D. N. Langenberg (Department of Physics

and Laboratory for Research on the Structure of Matter, University of

Pennsylvania, Philadelphia, PA 19174), Phys. Rev. B 1487-1522 (1 971 ).

An ac-Josephson-eff ect determination of e/h with significantly improved

accuracy is reported. The precision of the measurement is determined by

uncertainties associated with the comparison of a Josephson-device voltage

with the emf of an electrochemical-standard-cell voltage reference and is

o about 3 parts in 10°. This precision was made possible by use of Josephson

devices at voltages above 10 mV and design and construction of two special

voltage-comparator instruments. The fabrication and operation of the

Josephson devices and the design and performance of the voltage comparators

are discussed. The 3/1 0^ precision represents the precision with which a drift-free and readily reproducible Josephson voltage standard can be

realized in practice using the techniques developed for these experiments.

The accuracy of the final result is about 12 parts in 10^ and is determined

primarily by uncertainties associated with the stability of the local

electrochemical voltage standard and with establishment of the relationship

between the local volt and the volt maintained by the U.S. National Bureau of

Standards. Significant improvements in the maintenance of the local-voltage

standard which contributed to reduction of the final uncertainty to this

value are discussed. During the course of the experiments, the Josephson

frequency-voltage realtion was shown experimentally to be independent of

magnetic field, temperature, and Josephson-device bias voltage or induced

step number to within the accuracy of the final result. The final experimental result and its one-standard-deviation uncertainty are 2e/h

(483.593718+0.000060) MHz/yVNBS69 (0.12 ppm) referred to the volt as maintained by the U.S. National Bureau of Standards after January 1, 1969. This

result is in excellent agreement with the earlier, less accurate result of

Parker, Langenberg, Denenstein, and Taylor, which played an important role in

the 1969 adjustment of the fundamental physical constants by Taylor, Parker,

and Langenberg. It is in reasonable agreement with values recently reported

by several other groups. The significantly improved accuracy of the present

result makes possible a small improvement in the accuracy of the derived

value of the fine structure constant and clears the way for a larger improvement through more accurate determination of the proton gyromagnetic ratio.

233

VOLTAGE REFERENCE STANDARDS AND TRANSFER METHODS

A Sub-PPM Automated One-to-Ten Volt D.C. Measuring

SYSTEM

Bruce F. Field (1984) 237

An Automated Standard Cell Comparator Controlled

by an Desk Calculator: A Preliminary Report

Robert E. Kleimann and Woodward G. Eicke, Jr. (1976) 241

A Resistive Standard for Measuring Direct

Voltages to 10 KV

Clifton B. Childers and Ronald F. Dziuba (1976) 247

A High-Resolution Prototype System for Automatic

Measurement of Standard Cell Voltage

David W. Braudaway and Robert E. Kleimann (1975) 251

Standard Cell Enclosure with 20- uK Stability

Robert D. Cutkosky and Bruce F. Field (1974) 256

Standard Cell Calibration via Current Transfer

Edwin R. Williams, Paul T. Olsen and Bruce F. Field (1974) . . 260

The EMF-Temperature Coefficient of "Acid" Standard

Cells of the Saturated Cadmium Sulfate Type from

15 to 40 °C

Walter J. Hamer, Anna Skapars, and Bruce F. Field (1972) . . . 264

Thermodynamics of Standard Cells of the Saturated

Cadmium Sulfate Type

Walter J. Hamer (1971) 272

Designs for Surveillance of the Volt Maintained by

a Small Group of Saturated Standard Cells

W. G. Eicke and J. M. Cameron (1967) 293

Measurement Assurance Programs in a Field Environment

Woodward G. Eicke, Jr., Thomas F. Leedy, Brian R. Moore,

and Charles F. Brown (1982) 312

Transfer of the Unit of Voltage

Norman B. Belecki (1968) 321

Regional Maintenance of the Volt Using NBS Volt

Transfer Techniques

Woodward G. Eicke, Jr. and Laurel M. Auxier (1974) 327

A Coordinated System of Maintaining and Disseminating

the Volt

Robert M. Shaw (Abstract, 1980) 332

235
The Response of Standard Cells to Alternating Currents

in the Frequency Range 10-10^ Hz

G. J. Slogett (Abstract, 1977) 332

Automatic Intercomparison of Standard Cells

Andrew F. Dunn (Abstract, 1 97^ ) 332

Internal Comparison of a Large Group of Standard Cells

J. Wilbur-Ham (Abstract, 1973) 333

Automatic Measuring System for a Control of Standard

Cells

Hiroyaki Hirayama and Yasushi Murayam (Abstract, 1972) .... 333

Maintenance of a Laboratory Unit of Voltage

Andrew F. Dunn (Abstract, 1971) 333

Truly Transportable Standard Cell Air Bath

David W. Braudaway (Abstract, 1970) 33^

The Application of the Direct Current Comparator

to a Seven Decade Potentiometer

Malcolm P. MaeMartin and Norbert L. Kusters (Abstract, 1968) . 33^

The Construction and Characteristics of Standard Cells

George D. Vincent (Abstract, 1958) 335

236
A Sub-PPM Automated 1 — 10 Volt DC Measuring

System

BRUCE F. FIELD, member, ieee

Abstract— An automated measuring system has been developed for measuring arbitrary-voltage references in the range of 1-10 V with an

inherent measurement accuracy of ±0.22 ppm (3a). This paper dis- cusses the design and uncertainty analysis of the system and presents

data obtained on a Zener voltage reference.

I. Introduction

VOLTAGE standards based on solid-state devices are now

available with stabilities approaching 1 ppm/year [1] -[4]

.

These standards typically have their best accuracy at some

voltage other than the 1.01-V level of standard cells. In order

to separate the performance of the reference device from

ensuing buffer amplifiers, it is necessary to measure "peculiar"

Manuscript received August 20, 1984. This paper was supported in part by the Calibration Coordination Group of the U.S. Department of

Defense. The author is with the Electricity Division, National Bureau of Standards, Gaithersburg, MD 20899.

voltages, for example, 6.583 V. Typically, Kelvin-Varley

dividers are used to measure a wide range of voltages in terms of a standard cell. Accuracies better than one ppm are only

achieved by frequent, time consuming, manual calibrations of

the divider. As an alternative to the Kelvin-Varley method we have constructed a fully automated, self-calibrating measurement system to measure any arbitrary-voltage within the

voltage range of 1-10 V. The total measurement uncertainty

in comparing a voltage between 5 and 10V to a 1.0 18V stan- dard cell is ±0.22 ppm, three standard deviation estimate (3a).

II. System Description

The system to be described is essentially a potentiometric

technique which allows voltage measurements to be made

without loading the source being measured. Since the system was developed to measure Zener reference units and diodes, we will frequently refer to the voltage source being measured

as a Zener reference. Fig. 1 is a simplified diagram of the

237
LOW-PASS

FILTER

Fig. 1. Block diagram of the measuring system set up to compare a reference-under-test to the calibrated resistor string. Not shown are the crossbar selector switch and desktop computer used to control

the switch and DVM.

measuring system. It consists of 10 nominally equal 1000-S2

resistors (resistor string) driven by a 10.18-V battery-operated,

solid-state source (transfer reference) with good, short-term

stability. The measuring system is used in a two step process.

First, the voltage drop across each resistor is determined by

comparing it to a standard cell and measuring the microvoltlevel difference voltages with a high-accuracy digital voltmeter (DVM). Then the output voltage developed across N of the

resistors is compared to the voltage of the unknown Zener

reference using the same DVM. Following this, the voltage

drops across the resistors are again compared to the standard

cell voltage to estimate the drift in the resistor voltages during

the comparison. The voltage of the unknown Zener is obtained from the sum of the voltage drops across the TV resistors and

the digital voltmeter reading. A low-thermal, crossbar selector switch (not shown in Fig. 1) is used to connect the standard

cell and Zener references to the measurement system. Up to 60 voltage sources can be selected for measurement. The

DVM and the selector switch are both connected to a desktop

computer via the IEEE-488 bus so that the whole measure- ment process is completely controlled by the computer.

With this system the DVM is used to read only a fraction of

the unknown Zener reference voltage which reduces the contribution of the DVM uncertainty to the overall measurement

uncertainty. Thus

U = Vz - (TV) 1.01UDVM

where Udvm is the DVM uncertainty expressed as a percentage of reading, Vz

is the Zener voltage, TV is the number of resis- tors, and U is the final DVM uncertainty as a percentage of

the Zener voltage. For a worst case of Vz = 5.6 V, U - 0.09

(7DVM , and for Vz = 10 V, t/=0.02 UDVM . Therefore, if the DVM can be calibrated to an accuracy of 1 ppm the worst

case error contribution is 0.09 ppm. The gain of the DVM is calibrated during the measurements by periodically measuring

the 1.018 V developed across one or more of the resistors. The linearity of the DVM on the 1 0-V range can be checked by directly measuring the ten voltages, 1.018-10.18 V, devel- oped across the resistors.

III. Transfer Reference

The transfer reference is a commercial solid-state reference

that has been modified to produce 10.18 V rather than the 10 V normally supplied. When measurements are being taken,

the reference is disconnected from the ac power line and

operated from its internal batteries to eliminate coupling to the power line. When the reference is disconnected from the ac power the output voltage is observed to drift down approximately 0.2 ppm during the first hour, apparently caused by

temperature changes in the reference. This particular transfer reference has the power transformer in the same enclosure as the reference device and the latter is not temperature controlled. When power is removed from the transformer the

internal temperature typically drifts down by about 1°C during

the first hour. After a one hour "warm up" time the typical

output voltage drift is -0.018 ppm/h. Since the voltage drops

across the resistors (and hence the transfer reference), they are calibrated by standard cells before and after Zener measure- ments, only short-term stability is required of the transfer reference.

IV. Low-Pass Filter

The transfer reference was found to be sensitive to ac noise coupled into it from the output terminals. If any solid-state voltage reference is not properly isolated from external noise

sources, nonlinear circuit elements, including the Zener diode,

rectify the ac signals and produce a dc shift in the output

voltage [5] . Connecting the DVM directly to the transfer reference produced a dc shift in the output voltage of 0.2 ppm. This was determined by monitoring the reference out- put voltage with a passive measuring system while connecting

the DVM. The value of the dc shift was dependent on the

position of the DVM in the measuring circuit, and was ex- tremely reproducible from day-to-day. This produced a sys- tematic error in the calibration of the voltage drops of the 10

resistors. A three-stage RC PI low-pass filter with a cutoff frequency of 400 Hz was added between the output of the

transfer reference and the resistor string to reduce the ac noise. The filter forms part of a voltage divider (with the resistor string) so wire-wound resistors were used for stability and to reduce thermal EMF's. Installation of the filter reduced the worst case dc shift to 0.039 ppm. However, since similar shifts occur during determination of the resistor voltages and during

the unknown Zener measurements, the dc shift error is partially canceled. When necessary, filters are also added to the un- known Zener references under test.

V. Resistor String

The resistor string driven by the transfer reference consists of 10 nominally equal, stable, lOOO-H wire-wound resistors. We would have preferred to use lower-valued resistors to reduce measurement errors from leakage currents, but the transfer reference is limited to a load current of slightly more than

1 mA. The resistors are mounted in a temperature-controlled

air enclosure, with a simple solid-state sensor and dc amplifier

controller to maintain the short-term temperature variations to within ±0.01 °C.

VI. Crossbar Selector Switch

The crossbar selector switch, a modified commercial product

originally intended for telephone applications, permits up to

238
FIELD: SUB-PPM MEASURING SYSTEM

60 voltage sources to be compared to the voltage drops across the resistors or intercompared with each other and is a varia- tion of that described in [6] . The switch consists of a matrix

of switch contacts that are actuated by magnetically latched

solenoids to reduce the power dissipated in the switch. Gold

contacts on beryllium copper are used exclusively, and the switch is mounted in a temperature-lagged enclosure with all leads to the switch thermally connected to a large aluminum

plate. The crossbar switch is interfaced to the desktop computer via the IEEE bus. A universal coupler converts the 488

bus commands to parallel outputs which are boosted by

transistor-drivers to actuate the solenoids with the proper

timing sequences. Switch reliability has been excellent, no

switch problems having been encountered in two years of

operation. Thermal EMF's are periodically checked by short- ing the input leads and measuring the voltage with the DVM.

Unresolved thermal EMF's are less than 0.030 mV.

VII. Digital Voltmeter

A 6j digit DVM with 0.1-/uV resolution on the 0.1-V range

is used as the system voltmeter. The voltmeter is programmed

by the desktop computer via the IEEE-488 bus. All voltages

are measured by the DVM in both the forward and reverse direction to cancel offset voltages in the DVM. The polarity

reversal is done by the crossbar switch. The 0.1-V range is used when calibrating the resistor string voltages against stan- dard cells, and an integration time of 9 s for each polarity

yields a random measurement uncertainty of 0.034 uV (loO- For measurements of Zeners (as shown in Fig. 1) the voltmeter

is calibrated and used only on the one volt range.

VIII. Desktop Computer

The desktop computer used with the system is programmable

only in the Basic language, has 64 kbytes of main memory, and

has two 10 Mbyte disks. The large on-line storage capability

permits us to maintain a database on measurements made on all the references, examine data, plot control charts, and produce

reports as necessary. The system is fully automated with the computer controlling all aspects of the measurement process including operating the crossbar switch, reading the DVM,

supplying ac power to the references under test, and reading

the temperature of the standard cells. A number of programs have been developed to automate

completely the measurement process. These programs can be used individually or under the control of an interpretative program which calls the programs as overlays (because of

limited main memory) and stores the measurement data on a

disk. The major tasks include determining the voltage drops

across the resistors in terms of a group of standard cells, comparing the voltages across the resistors to a number of unknown

Zener references, and producing final calibration reports and

control charts on all the Zener references and standard cells. The interpreter program is very general, permitting any

sequence and/or number of measurements to be made based on a set of measurement commands previously stored on the

disk. If no changes are to be made to the previous days measurement sequence, performing a complete Zener reference

calibration consists of running the program at the start of each

day. However, daily changes in the measurement sequence can

easily be made, if desired. The standard cell voltages are compared to the voltages across the resistors using a redundant measurement design. The in- dividual cell and resistor voltages are then determined using

a least squares solution to the data. Zener references are identified in the system by a unique serial number with a translation table stored on the disk to equate the serial number

to the proper selector switch circuit. Multiple comparisons

are made between the Zener reference voltages and the voltages

across the resistors to permit estimation of the random measurement uncertainty.

The EMF values assigned the standard cells in terms of the NBS volt are maintained throughout all data files. If later calibrations indicate that small corrections to the cell voltages

are necessary, they can be applied to the final Zener voltage

results without recalculating all the intermediate data. The

latest corrections are maintained on the disk and are applied

automatically by the calibration report generator.

IX. Calibration and Sources of Uncertainty

By far the most critical component of the system is the

digital voltmeter. However, in this system the DVM offset and

gain are calibrated during the measurements. Provision is also made to calibrate automatically the linearity at 10 points on

the 10-V range by measuring the 10 voltages 1.018, 2.036, 10.18 V although periodic linearity measurements at 100

points using a Kelvin-Varley divider have shown that it is not

necessary to implement this feature. The system was designed

principally to measure Zener diode references in the range 5-10 V. Table I lists the sources of uncertainty in the measuring system extrapolated to the worst case unknown voltage

(5.5 V) in the 5-10 V range. The system may be used over

the range 1-5 V but with somewhat reduced accuracy when

the unknown voltage is not near a cardinal value (1 , 2, 3, • • •

, V). The uncertainties in Table I are discussed below. The DVM gain is measured during the course of the Zener measurements (usually four times) by reading the voltages across two or more individual resistors in the resistor string. From these measurements the gain error of the 1 V range can

be estimated. Two resistors are used for redundancy and to estimate the errors in measuring voltages above ground.

The DVM linearity is periodically checked on the 10-V range

using a calibrated, manual, 7-dial Kelvin-Varley divider. A

computer-aided system is used to log the data and typically 100 voltages are measured. Measurements performed when

the system was first set up indicated no substantial difference

in the linearity of the 10- and 1-V ranges. The linearity error on the particular DVM being used was typically 0.7 ppm at half-scale input. Measurements on three other meters of the same model have confirmed that the linearity error is typically

less than 1 ppm of full scale. The DVM leakage and bias currents and the standard cell leakage currents were measured directly with an electrometer and their effects on the measurement calculated. The ac

effects on the transfer reference were estimated by monitoring

changes in its voltage using a passive measurement circuit referenced to a standard cell while the DVM was measuring

239
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

TABLE I Uncertainties in the Automated Measuring System

Source of uncertainty

DVM gain uncertainty

DVM linearity uncertainty

DVM leakage/bias currents

Standard cell leakage currents

AC effects on transfer reference

Selector switch thermal emfs

Random uncertainty In calibrating

transfer reference

Random uncertainty in calibrating

unknown reference (within day)

RSS subtotal

1 std. dev. estimate

( ppm '/

0.020

0.062

0.023

0.006

0.023

0.013

0.013

0.075

Uncertainty in standard cell assignment 0.11

0.133

the resistor string voltages. The selector switch thermal EMF's

are periodically measured with the DVM by shorting the input

leads and measuring the residual voltages. The random uncertainties in calibrating the transfer reference and the unknown

references are estimated from the pooled standard deviations of the measurement designs.

Solid-state voltage references sent to NBS for calibration are now calibrated using this system. In this case an additional uncertainty must be included for the uncertainty of the EMF's

of the standard cells used to calibrate the system. The 0.11 ppm uncertainty listed in the table is not indicative of the best obtainable accuracy but is rather an estimate of the uncertainty obtained when the standard cells are calibrated in terms of the NBS volt only three times a year. We have recently installed a cable which will allow more frequent calibration of our cells but it is too early to estimate the improvement in the uncertainty using this method. As far as is known, all systematic errors listed in Table I are independent of each other. Therefore, they are combined in root-sum-square (RSS) fashion along with the random errors.

X. Results and Conclusions

Measurements of the output of a typical Zener reference are shown in Fig. 2. The reference exhibits a 1 ppm/year drift with a standard deviation of a single observation of 0.07 ppm. We have observed that most references measured by the system

showed an unexpectedly high correlation in their voltage fluc- tuations from day-to-day. This same correlation appeared

between references near a cardinal value (i.e., 1.018 V) where

the DVM reading is small, and 10 V where the DVM reading is

TIME (months)

Fig. 2. The measured voltage versus time of a commercial Zener refer- ence standard.

much larger. We concluded that the problem was related to measuring the temperature of our standard cells. Improve- ments were made to read the temperature automatically at the time of the measurement rather than 1 or 2 h before, and

the correlation was reduced. We have developed a fully automated measuring system

capable of calibrating an arbitrary-voltage reference with an

inherent accuracy of ±0.22 ppm (3a). It is based on the principle that a properly calibrated digital voltmeter can be used

as a transfer device to an accuracy of at least 1 ppm. Using

this system we have shown that some solid-state voltage references presently available have drift rates as low as 1 ppm/

year, nearly as good as standard cells. We have also found that some Zener references shift in voltage by as much as 2 ppm

when connected to a DVM and ac filtering must be used to reduce this offset. In addition, using the principles developed here, possibly in conjunction with fixed ratio voltage dividers, the voltage range can be extended up to 100-1000 V or down to 100 mV, albeit with reduced accuracy. It is also reasonable to use this system

as part of a self-calibrating dc reference in an automatic test equipment (ATE) system, or as a high-accuracy automated

DVM calibrator.

Acknowledgment

The author wishes to thank W. G. Eicke for his critical read- ing of the manuscript and many helpful suggestions.

References

[1] W. Murray and P. Dencher, "High-accuracy voltage transfer using a modified commercial solid-state reference," J. Phys. E.: ScL

Instrum., vol. 17, no. 5, pp. 354-356, 1984.

[2] P. J. Spreadbury and T. E. Everhart, "Ultra-stable portable voltage sources," in Conf. on Electron. Test and Measuring Instrum., IEE

Conf. Publ. 174, 1979, pp. 117-120.

[3] K. J. Koep, "The use of a solid state dc voltage transport standard

to transfer the dc volt in a regional standard cell MAP program,"

inProc. 1984 Meas. Science Conf., 1984, pp. 14 2- 146.

[4] L. Huntley, "The Fluke direct voltage maintenance program," in Proc. 1984 Meas. Science Conf., 1984, pp. 147-151.

[5] W. G. Eicke, "The operating characteristics of Zener reference diodes and their measurement," ISA Trans., vol. 3, no. 2, pp. 93-99, Apr. 1964.

[6] D. W. Braudaway and R. E. Kleimann, "A high-resolution proto- type system for automatic measurement of standard cell voltages," IEEE Trans. Instrum. Meas., vol. IM-23, pp. 282-286, 1974.

240
AN AUTOMATED STANDARD CELL COMPARATOR CONTROLLED JY A DESK CALCULATOR: A PRELIMINARY REPORT

Robert E. Kleimann and Woodward G. EJcke, Jr. Electricity Division

National Bureau of Standards

Washington, D.C. 20234

Abstract

A system is described with which measurements of standard cell differences can be made automatically to an accuracy of one part in 10^. The system is built

using all commercially available equipment and is designed to be a relatively inexpensive easy-to-use

instrument for the calibration laboratory. The system

is designed around a BASIC-programmable desk

calculator and MIDAS, an NBS-developed modular

interface, which together control the necessary

switching, stimuli and measurements for a set of standard cell difference measurements. The calculator

then processes the measurement data using a least-squares fit and prints a calibration report for

the cells. The design and construction of the system

is described along with an evaluation of the results

obtained with the system.

Introduction

Voltage measurements of standard cells are accomplished by connecting a standard cell of known

emf value in series opposition with a standard cell of unknown value. This is done for two main reasons.

First, by bucking out most of the voltage, the residual voltage need only be measured to a modest

accuracy in order to make a high accuracy

inter comparison of two cells. For example, a difference of 100 uV, measured hy a 0.1% instrument

would essentially resolve 0.1 pV or 1 part in 10^ in terms of the cell emf. Secondly, standard cell

performance is significantly degraded by current drawn

from or passed through them. A standard cell of typical voltage 1.018 V measured directly by an

instrument with 10 Mfl input impedance would, by Ohm's

law, result in a current of 101.8 nanoamperes. By

measuring a difference of, typically, 100 to 200 uV,

the current would be only 10 to 20 picoamperes.

Low-level, high-accuracy voltage measurements

require a consideration of some sources of error not

significant at higher levels or reduced accuracy. The

two major sources of error are thermal emfs and

leakage currents.

Thermal emfs are caused by temperature gradients

existing in the measurement circuit. * The Seebeck

effect is a thermal emf caused by a temperature

gradient across a junction of dissimilar conductors.

The thermal emf is a function of the types of conductors and is directly proportional to the temperature gradient. The Thompson effect is the thermal emf of a homogeneous conductor in a temperature gradient. This thermal emf is also a function of the type of conductor and is directly

proportional to the temperature gradient. Thermal

emfs in a circuit can be reduced to an acceptable

level by using the same or well-matched (in terms of thermoelectric effects) conductors throughout and

controlling temperature gradients.

Potential differences existing between conductors

in a circuit will cause a current to flow between

them, limited by the resistance. If these leakage currents are large enough, they will be a source of error in standard cell measurements because 1) the currents are drawn from the standard cells and 2) the measured emf of the cells will be less than the actual

emf by the value of the voltage drop across the internal resistance of the cells. Careful attention

to the amount and type of insulation surrounding the conductors and the use of shielded and/or guarded

conductors will reduce leakage currents to an

2 acceptable level.

Statistically designed sets of measurements on

3 standard cell groups have reduced the number of measurements required to characterize standard cells

while at the same time yielding as results 1) the calculated emf of each cell in the group 2) an estimation of systematic effects caused by thermal

emfs and other offsets (the left-right component) 3) the precision of the measurements by the estimation of

3 4 the standard deviation of a single observation. ' Computer programs for the least-squares fit of measurement data which give the results listed above

have been implemented in both BASIC and FORTRAN. 5 '*'

The automated standard cell comparator described

below was designed to be a relatively inexpensive,

easy-to-use instrument for the calibration laboratory.

It was designed to have an accuracy of at least one part in 10^ and uses commercially available equipment

as much as possible.

This paper is a contribution of the Wationai Bureau of Standards, not subject to copyright.

This work was supported by the Calibration Coordination Group of the Department of Defense.

241
Measuring Circuit Hardware

Standard Cell Selection and Circuit Configuration

The selection of a pair of standard cells whose

difference is to be measured and the connection of the measuring instruments in the desired

configuration is accomplished by a coordinate-type

electromagnetic crossbar switch. This switch has 100-input channels of 6-levels each and 10-output

channels of 6-levels each. Each output channel can select any of 10-input channels. Output channels 1,

3, 5 and 7 are wired in parallel to form a left system bus and output channels 2, 4, 6 and 8 are wired in parallel to form a right system bus. Two standard cells are each connected to two input

channels, one channel selectable for the left system

bus and one channel selectable for the right system

bus. (See Figure 1.) Since only 4-levels are needed

for two standard cells, the remaining 2-levels are wired to be indicators of the selected cells. Each

system bus line, then, by closing one crosspoint,

will have two cells selected. The ninth output

channel has the two system bus lines as its ' inputs.

By closing one crosspoint on this channel, one cell

on the left system bus will be placed in series

opposition with one cell on the right system bus with

either negative sides common or positive sides

common. Thus, any pair of eighty standard cells can

be connected, with either positive or negative

common, through only two sets of contacts. The tenth

output channel is used for selecting a measurement

circuit configuration.

To reduce thermal emfs to a minimum, temperature

gradients mast be carefully controlled. The switch

is mounted inside two nested aluminum boxes with a layer of foam insulation between. This makes for a heavier than usual lagging of the switch environment

from the laboratory. Magnetic latching actuators are employed on the output channels so that no power is needed to hold the actuators in the operate mode;

pulses of 40 - 50 msec duration are sufficient to

operate or release the actuators. Essentially, there are no heat sources inside the enclosure. To reduce

heat flow along the cell leads and into the switch

contacts, a thermal shunt method is employed. Copper

wires from the switch terminals are brought through a massive aluminum plate by passing each through a drilled nylon screw. Each wire is then clamped

between two copper washers electrically, but not thermally, insulated from the aluminum plate by a beryllium oxide washer. Leads from standard cells

are also clamped between the two copper washers to complete the connection to the switch.

Voltage Divider

A voltage divider network used in the measuring

circuit (Figure 2) to provide a reverse emf is constructed by the series connection of a precision

10 Kfi card resistor and a precision 4-terminal in card resistor. The two resistors are mounted inside

an aluminum box and are connected to the switch

through the same type of thermal tie-downs described

above

Detector

The detector is a commercial low-level digital

voltmeter that is capable of resolving 0.01 uV and has a full-scale range of 10 yV on the lowest range.

The steady-state input impedance of this instrument

is somewhat greater than 10 tin. The transient

impedance, however, starts at about 8 Kfl but rises

rapidly to its final (steady-state) value. Standard

cells are not adversely affected by the small amount

of energy drawn from them in the transient state.

The detector is connected to the switch using thermal

tie-downs and has a guarded input. Since the detector is used to measure voltages of 1 uV or less, accuracy requirements are only modest. It is possible to calibrate the detector as required using

the automated system.

CELL 1 CELL 41 CELL 2 CELL 40 CELL 80

LEFT

SYSTEM

BUS

RIGHT

SYSTEM

BUS

FIGURE 1. STANDARD CELL SELECTION

242
10 K

LOW-LEVEL

DETECTOR

PROGRAMMABLE

VOLTAGE

SOURCE

FIGURE 2. MEASUREMENT CIRCUIT

MIDAS

CROSSBAR

SWITCH

DIGITAL

VOLTMETER

STANDARD

CELL

GROUPS

JL

10000:1

VOLTAGE

DIVIDER

PROGRAMMABLE

VOLTAGE

SOURCE

CONTROL LINES

DATA LINES

LOW-LEVEL SIGNAL LINES

FIGURE 3. BLOCK DIAGRAM OF AUTOMATED STANDARD CELL COMPARATOR

243
Progrnmnab le Voltage Source

A programmable voltage source is used in the measuring circuit to provide a precision reverse voltage across the IP: resistor of the voltage divider.

It has 5-progranmable BCD digits giving a maxinun of + 16.6665 volts at 100 nA maximum current. The accuracy is rated at + (0.01^ + 100 yV) and stability

.it + (10 ppm + 20 mV) . The programmable voltage

source has a guarded output and is connected to the switch using thermal tie-downs. The combination of a 10,000:1 voltage divider and a 16 volt maximum voltage

source limits the maximum cell difference measurable

to 1600 yV.

Digital Voltmeter

The digital voltmeter is used in the measuring

circuit in two ways: 1) to roughly measure large

unknown standard cell differences 2) to check the calibration of the programmable voltage source. It is

a 5-1/2 digit instrument with lowest range full-scale

reading of 0.1 V and resolution of 1 uV. It has a rated input impedance of greater than 1000 Mft and a guarded input. It, too, is connected to the switch

using thermal tie-downs. The guards on all instruments are tied together at the low side of the detector to eliminate ground loops.

Calculator and Digital Circuitry

The programmable desk calculator controls all standard cell and instrument switching, programs

and/or reads the instruments, processes the measurement data and prints the results. It is programmable in BASIC, the easiest and most widely

known high level language. It has matrix

manipulation, string variable and extended

input/output capability, 8 K words of read-write

memory, line printer, external tape cassette drive and serial I/O (EIA RS-232) interface.

The calculator is interfaced to the switching and

instruments through MIDAS^, an HBS-developed,

commercially available, modular interface. (See

Figure 3.) All instruments and the crossbar switch

are interfaced to individual MIDAS modules. The modules communicate bi-directionally with the MIDAS

controller using parallel ASCII and the MIDAS controller communicates bi-directionally with the calculator using serial ASCII. Thus only one I/O

slot in the calculator is used to interface to all the various equipment needed for the standard cell

comparator.

The major difficulty in interfacing the calculator to MIDAS is a timing problem. Since the calculator is much slower (100 - 1000 times) than a computer, MIDAS responds to a "SEND DATA" command

before the calculator is ready to receive the data.

At a data rate of 600 baud, a 12-bit long shift

register, clocked by the internal MIDAS clock delays

the data sufficiently for the calculator to be ready

to receive it. Also, the echo-back feature of the calculator's serial interface must be disabled so that

M.IDAS will not mistake its own data for control

commands.

The interfaces from MIDAS to the instruments are straightforward since the detector, programmable

voltage source and digital voltmeter have BCD inputs

or outputs and MIDAS has the necessary BCD modules.

A MIDAS crossbar controller module is usr>d I'o control the crossbar switch. The nodule contains the necessary logic, timing and transistor drivers to close any crosspoint selected. Power for the switch

is from an external power supply.

Measurement Design

When two standard cells of emf values E and E

? are connected in series opposition across a suitable

measuring instrument, the measured difference, ->E should theoretically be E

1 - E^ . Realistically,

however, there may be thermal emfs, instrument zero offsets or other spurious enfs in the measuring

circuit. The actual measured difference would be

AE = E. - E- + E d)

1 2 p where E is the sun of all spurious emfs in the

P circuit. (See Figure 4.) If a second measurement,

(Figure 5) with the opposite cell leads common, is taken and assuming the spurious emfs to remain

constant, the measured difference would be

AE' = -E. + E_ + E (2)

1 2 p An estimate of E is obtained from the average of

P the two measurements,

E = 1/2(AE + AE') (3) P and an estimate of - E^ , free of E

p is obtained

from the difference average of the two measurements

E - E

2 = 1/2(AE - AE') CO

The automated standard cell comparator uses this

technique, taking two measurements in opposite

polarities, for each cell difference to be obtained.

MEASURING

INSTRUMENT

+

FIGURE 4. FIRST MEASUREMENT

244
MEASURING

INSTRUMENT

+

«-AE- 1

E

E. EFIGURE 5. SECOND MEASUREMENT

11) Disconnect instruments.

12) Calculate cell difference from data obtained.

13) Connect same pair of standard cells in series

opposition with positive sides common. Wait for settling.

14) Connect voltage divider, detector and programmable voltage source.

15) Calculate a setting for the programmable voltage

source as the same magnitude of the last setting used

in the first cell difference measurement but with the opposite polarity.

16) Repeat steps 5) through 12).

17) Calculate E12- E (free of E ) from the two

P calculated cell differences.

The above procedure can be simplified if approximate cell values are known. For example, the detector has a 100% overrange; therefore, if the cell

difference is known to be less than + 20 yV, steps 2) and 3), reading the cell difference with the digital

voltmeter could be eliminated. Also, if the approximate cell difference is known, the first

setting of the programmable voltage source could be

calculated based on that known value. Only if the detector indicated an overrange condition would the digital voltmeter need be used to read the cell difference.

A typical programmed procedure for measuring a standard cell difference is as follows:

1) Connect the selected pair of standard cells in series opposition with negative sides common and wait

for settling.

2) Connect digital voltmeter across cells and read digital voltmeter for cell difference.

3) Disconnect digital voltmeter, connect voltage

divider, detector and programmable voltage source.

(Figure 2)

A) Calculate a setting for the programmable voltage

source so that the voltage across the In arm of the divider will equal the cell difference measured by the digital voltmeter.

5) Set programmable voltage source to calculated value

and wait for settling.

6) Read remaining difference with detector.

7) If difference is greater than + 1 uV, calculate a new setting for the programmable voltage source and go back to step 5; otherwise, continue.

8) Read remaining difference with detector ten times

and calculate mean and standard deviation of the ten

readings.

9) If the standard deviation exceeds the accepted

standard deviation for a single observation then the cells have probably not yet stabilized and step 8 should be repeated; otherwise, continue.

10) Disconnect standard cells; disconnect all instruments; then connect digital voltmeter across programmable voltage source and read digital

voltmeter.

measurement known group calibrated, measurements

If the programmable voltage source is sufficiently well-calibrated, it would not be necessary to connect the digital voltmeter across it to measure the output; the output could be assumed

equal to the known setting.

The calibration of a group of standard cells is usually accomplished by making a set of difference

measurements as described above. Each difference

involves one cell from a standard or

and one cell from the group to be

the unknown group. After all necessary

have been made and the data stored in the calculator memory, the calculator processes the data, doing a least-squares fit by means of a matrix

inversion technique to obtain emf values for each

cell in both groups. These results, along with the other statistics discussed in the introduction are printed as an NBS calibration report. The results

can also, if desired, be stored on a data tape for future use or analysis. Alternatively, the original measurement data can be stored on a tape for later

processing and another set of difference measurements

initiated. In either case, data needed for bookkeeping purposes; e.g., the date, cell serial

numbers, run number, etc., can be entered via the

calculator keyboard or an input data tape.

Results

The automated standard cell comparator has performed very well as of this writing. The time

required for a complete set of 16 measurements to be

made, calculations performed and a report printed is about 12 minutes. This compares favorably with the

time of 18 - 20 minutes required for only making the

same measurements manually. The left-right

component, a measure of the systematic effects caused

by constant circuit thermal emfs and other offsets, averages between 20 and 30 nanovolts. The observed

standard deviation, a measure of the precision of the

245
calibracion, has varied between 10 and 50 nanovolts,

apparently depending on the quality of the standard

cells. Measurement designs done manually for routine

calibration have an average left-right component of about 20 nanovolts and a pooled observed standard

deviation of about 25 nanovolts with some as low as 5 nanovolts. In instances where an automated design has been followed by the same design taken manually, or vice versa, the individual cell erafs have agreed to within 0.03 uV and the individual cell difference measurements have agreed to within 0.10 \iV . A series

of 29 designs involving the same cells, taken over a four-hour period has shown that the automated system

is treating the standard cells involved in a measurement very gently. Hie standard deviation of

the 29 calculated cell emfs was no greater than 0.014

uV for any of the standard cells.

6. B. F. Field, W. G. Eicke, Jr. and R. E. Kleimann,

Unpublished computer programs for reducing standard cell data.

7. Charles H. Popenoe and Mack S. Campbell, MIDAS : Modular Interactive Data Acquisition System - De- scription and Specification National Bureau of Standards Technical Note 790, U. S. Government Printing Office, (August, 1973).

Conclusion

The use of a programmable desk calculator as the controller of the system rather than a minicomputer

has enabled the design goals of "relatively

inexpensive" and "easy-to-use" to be met. The total cost of all macerials should not be greater than

$30,000. Software costs are low since there is no operating system to deal with; a BASIC interpreter is always available. There are no assembly language

drivers to write; functions or subroutines can be

easily written to output the necessary MIDAS commands.

Since standard cell measurements are inherently slow,

the speed obtainable with a minicomputer is not needed. The system is entirely transportable; all equipment, including the calculator, is mounted in a 6-1/2 foot high rack.

At this point, temperature measurements of standard cell enclosures must be made manually and the data entered separately. Modifications to some enclosures are being considered so that the system

will be able to measure temperatures automatically.

The system can easily be used as a data

acquisition system. Its versatility in this regard is limited only by the ability of interfacing equipment

to MIDAS. An extension of the system, adding another

crossbar switch and a suitable programmable power

supply, will enable automated precision resistance measurements to be done.

References

1. Donald I. Finch, General Principles of Thermoelectric Thermometry , "Temperature: Its Measurement and Control in Science and Industry", Volume

3, Part 2. Reinhold Publishing Corporation, New

York, New York, 1962.

2. Norman B. Belecki, A Guarded Shielded Standard

Cell Measuring System , ISA Preprint 706-70, (Oct.,

1970)

3. W. G. Eicke, Jr. and J. M. Cameron, Designs for Intercompar lng Small Groups of Saturated Standard Cells , National Bureau of Standards Technical

Note 430, U.S. Government Printing Office, (April,

1967) .

4. W. G. Eicke, Jr., The Volt , Yesterday , Today and Tomorrow , ISA Reprint 71-703, (Oct., 1971).

5. J. W. Linnik, Method of Least Squares and Principles of the Theory of Observations Translated

from the Russian by Regina C. Elandt. Pergamon

Press, New York, 1961.

246
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-25, NO. 4, DECEMBER 1976 A Resistive Ratio Standard for Measuring Direct Voltages

to 10 kV

CLIFTON B. CHILDERS, RONALD F. DZIUBA, MEMBER, IEEE, AND LAI H. LEE

Abstract—A highly accurate guarded voltage-ratio standard has been developed for measuring direct voltages from the range of a standard cell to 10 kV. The ratio standard has a resolution capability of 0.1 ppm for ratios of 1:1 to 10 000:1 with an estimated uncertainty of 0.2 ppm. It is designed for operation at a rating of 2 kfi/V and consists of an adjustable reference section in series with

three resistance groups each containing nine nominally equal

sections. The resistance ratios are determined by a self-calibration technique using a 1:1 bridge. A series-parallel mode of calibration provides an additional check on the accuracies of the ratios. The

standard is housed in a temperature-controlled oil bath whose oil

is filtered to remove moisture and contaminants.

I. Introduction

THE MEASUREMENT of direct voltages higher

than that of a standard cell is achieved by special re- sistive ratio standards having a limited number of discrete

ratios for operation at convenient voltage levels. At NBS,

a voltage-ratio standard [1] having ratio accuracies of a few

parts in 107 is used to calibrate commercial units up to the 1-kV level. Another NBS standard [2], designed for use to 100 kV, provides ratio accuracies of a few parts in 105

. There have been very few reports [3] of voltage dividers designed specifically for operation at the 10-kV level. This paper describes a highly accurate, guarded volt- age-ratio standard which has been developed for measuring direct voltages from the range of a standard cell to 10 kV: The ratio standard has a resolution capability of 0.1 ppm for ratios of 1:1 to 10 000:1 with an estimated uncertainty of 0.2 ppm. It will provide increased accuracies in the measurement of direct voltages from the 1 to 10 kV

range not possible with present NBS standards. An immediate application of this ratio standard is in the absolute

volt determination at NBS. In this experiment, the force between the electrodes of an electrometer with high voltage

applied will be compared to the force acting on a known

mass. Using the ratio standard, the applied voltages in the range of 4 to 9 kV will be measured in terms of one or more

standard cells. No auxiliary potentiometer or null bal- ancing circuits are necessary for these measurements be- cause of the fine resolution capability of the ratio stan- dard.

II. Design of Ratio Standard

The basic circuit of the ratio standard without the guarding network is shown in Fig. 1. It consists of an adjustable reference section followed by three resistance

Manuscript received June 29, 1976 The authors are with the Electricity Division, National Bureau of Standards, Washington, DC 20234.

groups A, B, and C each containing nine nominally equal

sections of 20 kQ, 200 kfi, and 2 MSI, respectively. The

reference section has a nominal value of 20 kfi and is comprised of an 8-decade Kelvin-Varley (K-V) divider of

100-kft input resistance shunted by a 25-kfi section. With

the input of the reference section as a base, 27 fixed ratios are available, extending over three decades to a maximum

of 1000:1. Since the reference section can resolve 10-8 of

input, the ratio standard has at least a resolution capability of 0.1 ppm for ratios of 1:1 to 10 000:1.

A. Main Circuit The main circuit of the ratio standard is constructed

from carefully selected card-wound resistors. Each section of groups A and B is comprised of two resistors, while ten

resistors are used for each section in group C. Each section of the ratio standard exhibits a temperature coefficient of

resistance (TCR) with absolute value less than 0.5 ppm/°C

by proper selection and matching of positive and negative

coefficients of individual resistors. Initially, the resistance

ratios were adjusted to within 2 ppm of nominal by special

trimming techniques. At rated voltage, the maximum

power dissipation for the resistors (6.4 cm X 2.5 cm) in group C is 1.6 mW/cm2

. At this power level, it is estimated

that the maximum temperature rise of the resistors in their

oil bath will be less than 0.2°C.

B. K-V Divider

The card-wound resistors of the first two decades of the K-V divider are connected to plug-type coaxial terminals. This method of mounting simplifies the guarding of each

resistor and provides for easy calibration of these decades. Each section contains its own trimming network that

provides adjustment over a 30 ppm range with better than

0.1 ppm resolution. The TCR's of the 10- and 2-kft sections are matched to within 0.1 ppm/°C and 0.5 ppm/°C, re- spectively. A commercial divider of 1 ppm accuracy completes the last six decades of the K-V divider. The K-V

divider is housed in its own copper enclosure which is immersed in the oil bath.

C. Guard Circuit

Electrically, the guarding network is the same as the main circuit. It maintains the proper potential on the

shields of the coaxial terminals used to mount the resis- tance sections, including the first two decades of the K-V

divider, to the cross-linked polystyrene top piece. The

switch housings of the commercial K-V divider are also

247
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1976

100B lOOt TOOK

Fig. 1. Circuit diagram (except for guard network) of 10-kV ratio standard.

connected to the guard circuit. In the oil, the main resistors are uniquely guarded and shielded by sandwiching each

resistor between two conducting plates of the same size as the resistors and 2.5 cm apart as shown in Fig. 2. With the

plates maintained at the proper potential, leakage currents are directed from shield to shield and confined to the guarding network.

D. Oil Bath

The oil bath in which the ratio standard is immersed (75 cm diameter X 40 cm deep) has copper walls about 6 mm

thick. Four Peltier units spaced at 90° around the enclo- sure are operated to supply or extract heat as the bath

temperature varies around its set point. Oil flow is directed upward across the resistance units by a central impeller

located in the bottom of the tank and operated through a mercury-seal bearing. The return flow of oil is directed

along the copper enclosure wall by an acrylic baffle; and

a platinum resistor, distributed throughout the annulus

between baffle and copper wall, acts as temperature sensor. This sensor and a reference resistor located in the bath

form two arms of a bridge whose output is amplified to supply the Peltier units. With 3 in of foam insulation around the bath, temperature control is better than 0.01 °C

when the divider is operated at 10 kV with a power dissi- pation of 10 W. The bath oil is filtered to remove moisture

and contaminants. The conductivity of the oil is also monitored to determine its quality.

III. Calibration of Ratio Standard

The ratio standard is designed to have negligible errors that result from corona currents or leakage currents through nearby insulation. During operation of the stan- dard, the input and output currents are equal. Consequently, its calibration or the determination of the values

of its ratios reduces to measuring differences of resistance and changes of resistance under different load conditions. The ratio of the input voltage V, to the voltage across the

reference section Vr is given by

Vi/VR =N = N'[l + ck ] where N' is the nominal ratio and cn is the correction in proportional parts. The output voltage Va of the reference

section can be expressed as VG = kVR [1 + c„]

Fig. 2. Guarding of main resistors. (1) Main resistor. (2) Guard resistor.

(3) Guard plates. (4) Teflon terminal. (5) Insulating rod.

where k is the dial setting of the K-V divider and Ck is the

correction to the dial setting. Therefore, for a particular

ratio and dial setting,

Vi/V0 = N/k [l + cn -ck ]. Two methods have been used to determine the Cn 's of the

ratio standard. One is a stepup method whereby a value

for every ratio is obtained. The other is a series-parallel

method which serves as a check for the 10:1, 100:1, and

1000:1 ratios and provides information concerning the load corrections of the standard.

A. Stepup Method

In the series mode of the ratio standard as shown in Fig.

1, the resistance of each section is nominally equal to either the resistance of the preceding section or to the summation

of all preceding sections. Because of this design, the values

of cn can be determined by measuring small differences of resistance of nominally-equal sections. The resistance comparisons are performed using a guarded 1:1 bridge as schematically shown in Fig. 3. Ratio arms A and B of 1 kfi each comprise a Direct-Reading Ratio Set (DRRS) having

a least count of 0.1 ppm. (Interpolation of the detector unbalance extends the resolution of the system.) The R

arm indicates the section of the standard under test. Resistor S, depending on the section under test, is nominally

equal to 20 kQ, 200 kfi, or 2 MQ. Resistors a, b, r, and s form

the guard network which drives the shields of all leads at the proper potential to eliminate errors caused by leakage

248
CHILDERS et al.: RESISTIVE RATIO STANDARD

Fig. 3. Bridge circuit for calibrating ratio standard.

Fig. 4. Series-parallel configuration of a resistance group.

currents. The completely shielded system is immune from

external electrostatic interference. A set of 10 measurement points is obtained for each re- sistance group in order to compute the values for cn . Equations for computing these corrections are given in [1]. The measurement of the first section of a group is repeated

at the end of the set to check on the stability of resistor S. The total set of 33 measurement points takes approximately one hour to complete. Measurements are per- formed at rated voltage, i.e., 2 kft/V. The detector consists of a photoelectric galvanometer system operating at a

sensitivity of 20 nV/mm. Noise in the bridge network limits the useful detector sensitivity to approximately 25 nV or

0.05 ppm.

B. Series-Parallel Method

In addition to the series mode, the nine sections of a given resistance group can be connected into a parallel

configuration. A pair of bifurcated leads, as shown in Fig.

4, are used for the series-to-parallel conversion and sub- sequent connection to the DRRS bridge. The parallel re- sistance of groups A, B, and C are nominally 20 kft, 200 kO,

and 2 MQ, respectively. Each group in the parallel configuration is compared to all of the series-connected sec- tions that precede it. A set of six measurement points are obtained for computing the corrections to the 10:1, 100:1, and 1000:1 ratios. Measurements of the parallel-connected

sections are performed at % rated voltage or % the power

level. Consequently, differences between this method and

the step-up method will provide data on the load corrections of the standard.

C. K-V Divider

As previously mentioned, all 22 sections of the first two

decades of the K-V divider are accessible at the top panel.

The linearity of each decade is determined by intercom

paring its sections using a bridge circuit similar to the one shown in Fig. 3. To eliminate the need for applying lin- earity corrections, the trimming networks of each section are adjusted during the measurements to provide a lin- earity of better than 0.1 ppm. The linearity of the commercial divider, which completes the last six decades of the K-V divider, can be checked by standard calibration techniques [4], [5]. A determination of the end corrections of the K-V di- vider due to lead and contact resistances is necessary to complete the calibration. This is accomplished by energizing the divider at some convenient voltage and then

measuring the voltage between the output terminal and

each low terminal with all the dials set at zero; and, measuring the voltage between the output terminal and the high input terminal with all dials set at full scale. The

uncertainties that may arise from self-heating are negligible, since maximum power dissipation in the divider is

1 mW and the TCR's are small.

IV. Experimental Results

A number of calibrations have been performed since the

ratio standard was immersed in its oil bath in February

1976. An analysis of the data accumulated during the fol- lowing three months indicated that the values of ratio were

correlated with changes in the relative humidity of the

laboratory. Differences of ratio as large as 0.5 ppm were

observed over a 24-h period when the ambient relative humidity changed from 30 to 60 percent. It should be noted

that the top panel of the standard does not make a tight

seal with the oil bath and it is very probable that the air space above the bath reflects ambient conditions in the

laboratory. In addition, with the oil being stirred, the re- sistance changes with respect to changes in ambient humidity occur in a relatively short period of time. To correct the situation, a small amount of filtered and dessicantdried air under pressure is being continually forced above

the oil bath. Table I compares the results between the step- up

method and the series-parallel method for a set of 10 complete calibrations taken over a 10-day period started two days after the humidity control system was installed. For comparison, only the 10:1, 100:1, and 1000:1 results are given for the step-up method. Since the measurements

were performed, a systematic error of +0.04 ppm was

discovered when measuring the 20-kl2 reference section during the series-parallel calibration. This reduces the

differences between the two methods to approximately

0.01 ppm. The additional 0.11 ppm difference for the

1000:1 ratio results from loading of the 2-Mfi sections. 249
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1976

TABLE I

Compari on of Step-up (S-U) and Series-Parallel (S-P) Calibratio n Methods

Percent Mean Standard

Ratio He t hod Rated Deviation of

Voltap.e (ppm) Mean (ppm) 10] S-U 100 O.ofc 0.03 S-P 33 1/3 •0.89 0.02

100:] s-u 100 1.25 0.03 S-P 33 1/3 +1.30 0.03

1000:1 s-u 100 -0.83 O.Oli S-P 33 1/3 -0.66 0.06

TABLE II

Estimates of the Effects of Possible Systematic Uncertai nties

S.y sterna t ic Uncertainties

Estimate*

(ppm)

Resistance Comparisons 0.07

K-V Divider Calibration 0.05

Corona Currents 0.03

Leakage Currents 0.01

Load Coefficient o.ofc

Interconnections 0.02

Short-term Stability of Resistors 0.03

direct sum: 0.25

rss

o.n

These values are broad subjective es statistical 2o - limits.t i mates intended to be comparable with

Estimates of the effects of possible systematic uncertainties are itemized in Table II. The uncertainties of the

resistance comparisons and K-V divider calibration primarily reflect the uncertainty in the calibration of the DRRS and the stability of resistor S of Fig. 3. No visual corona was observed when the ratio standard was energized to 10 kV in air. Since the dielectric strength of the oil

is more than 20 times that of the air and the onset of visual corona is within 10 percent of the threshold voltage of

significant dark (or precorona) currents, it is estimated

that these uncertainties are within 0.03 ppm. The insula- tion resistance between the main and guard circuits is >1013 U. This resistance is effectively increased by 104 with

the guard ratios trimmed to 100 ppm of nominal. Thus

leakage uncertainties are considered to be negligible. At

the 20-kS2 level, interconnection uncertainties due to leads, terminations and cross resistances are within 0.02 ppm.

The load coefficient uncertainty only enters in when the standard is operated at a voltage different from its cali- bration voltages. A 0.03 ppm uncertainty is estimated for the effects of short-term variations of temperature and

humidity on the resistance elements. The total rss sys- tematic uncertainty is estimated to be 0.11 ppm.

The imprecision and the estimated systematic uncertainty of the measurement process appear to be of the same

magnitude. The overall uncertainty of 0.2 ppm is based on

2a and the total rss systematic uncertainty. The relatively

large standard deviation of 0.05 ppm is due to electrostatic noise in the system resulting from low humidity conditions and the stirring of the oil.

V. Conclusion

Experimental study of this 10-kV ratio standard indi- cates an uncertainty of its ratio corrections of 0.2 ppm. It will be used for the measurement of high voltage in the determination of the absolute volt. The development of

this standard can also lead to a more accurate assignment

of ratios and direct voltages to 10 kV. Investigation of the

stability of the standard will continue.

Acknowledgment

The authors wish to thank F. K. Harris for contributing

ideas regarding the design and construction of the stan- dard.

References

(1) R. F. Dziuba and B. L. Dunfee, "Resistive voltage-ratio standard and

measuring circuit," IEEE Trans. Inst rum. Measure., vol. IM-19, p. 266, 1970.

[2) J. H. Park, "Special shielded resistor for high-voltage dc measurements," J. Res. Nat. Bur. Stand., vol. 66C, p. 19, 1962.

[3] H. Hirayama, M. Kobayashi, K. Murakami, and T. Kato, "10-kV

high-accuracy dc voltage divider," IEEE Trans, lnstrum. Measure.,

vol. IM-23, p. 314, 1974.

|4| A. F. Dunn, "Calibration of a Kelvin-Varley voltage divider," IEEE

Trans. lnstrum. Meas., vol. IM-13, p. 129, 1964.

[5] D. G. M. Buist, "Calibration of a Kelvin-Varley voltage divider,"

A. W.A. Technical Rev., vol. 14, p. 323, 1971.

250
Reprinted by permission from IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT

Vol. IM-23, No. 4, December 1974, pp. 282-286 Copyright 1975 by the Institute of Electrical and Electronics Engineers, Inc. PRINTED IN THE U.S.A.

A High-Resolution Prototype System for Automatic

Measurement of Standard Cell Voltage

DAVID W. BRAUDAWAY, member, ieee, and ROBERT E. KLEIMANN

Abstract—The conceptual requirements, the design, and the initial performance of the prototype switching for connecting standard cells

to an automatic measurement system are described. Features of the

design include random selection of two cells at a time, inversion of connection polarity on command, modular construction for expansion, and less than 10-nV residual uncompensated error voltage. Also described briefly are controllers for manual operation of the

switches and the rudimentary high-resolution digital potentiometer used to complete the measurement system. Results of tests of the

switching and observations of the prototype system performance are

presented.

I. INTRODUCTION

SINCE their development late in the nineteenth century

by Weston [1], [2], standard cells have been employed in maintaining a standard of voltage throughout

the world at the highest attainable levels of accuracy. Testing of these cells has long been the domain of highly

skilled and dedicated personnel whose efforts brought the

world to the level of the "part-in-a-million" volt. The sheer

Manuscript received July 3, 1974; revised August 28, 1974. This

paper is a contribution of the National Bureau of Standards and

not subject to copyright. D. W. Braudaway is with Sandia Laboratories, Albuquerque,

N. Mex., and served as Guest Worker at the National Bureau of Standards, Washington, D. C, during this research. R. E. Kleimann is with the National Bureau of Standards, Washington, D. C.

burden of numbers of cells to be tested in recent years, the

desirability of employing advanced test programs with

computer aided analysis of the data [3], and the recent development of volt maintenance using the ac Josephson

effect [4] have led to the need for automating standard

cell measurements. Efforts toward this goal have ranged

from the simple obvious use of a high-impedance digital

voltmeter to elaborate systems employing scanning and

achieving resolution on automatic tests of 1 part in 107 [5]-[7].

Recently, a project was initiated at the National Bureau

of Standards, Gaithersburg, Md., the long-range objective

of which is the automation of all measurements necessary

to characterize the performance of both standard cells and

their enclosures. Levels of precision and accuracy sought

are commensurate with the highest requirement likely to

arise in the foreseeable future. The purpose of this paper

is to describe the development of the prototype system

and to describe the initial experience with the system in testing and use. The goal sought for performance of the

prototype system is that the error assignable to it be less than 0.01 mV, or 1 part in 108 of standard cell voltage.

II. SWITCHING REQUIREMENTS

Required for any highly precise automated measuring

system is the switching necessary for scanning or inter- connecting the various parts of the total system load. For

251
BRAL'DAWAY AND KLtlMANN: HIGH-RKSOLUTION PROTOTYPE SYSTKM

a system capable of meeting future needs, not only must

a high order of electrical performance be achieved, but also a sufficient flexibility must be provided to satisfy demands

imposed by test philosophies yet in the throes of creation. To meet these performance criteria, a list of requirements was established for the prototype switching which are enumerated as follows along with the associated reasoning.

1) Random selection of cells, two at a time, with either a normal or an inverted polarity was required.

Random selection insures maximum flexibility. Pairing

of cells permits intercomparison by difference measure- ment and produces a left-right sense for optimum use of

statistically derived measurement patterns [8]. Polarity

inversion, suitably interlocked for protection of the cells, permits a cancellation of nearly all unwanted parasitic

voltages.

2) Modular construction with a provision for local use

of the cells connected to any module without disrupting the system was required.

Modular arrangement, besides permitting the switching

function to be accomplished near the cells involved in the

test, also permits a simple change of the system size. Cells may be conveniently addressed by their location within the module complement, the module identifier, the output

line identifier, and the polarity identifier. This pattern of operation is readily interfaced to computer control, and

a simple highly reliable interlock is possible to preclude

accidental switching of cells in parallel or with opposite

polarity.

3) Simplified connection within the system using only

two pairs of guardable coaxial cables was required. Use of only two lines to interconnect modules and

measuring apparatus permits simplified system wiring and

minimizes loading of cells by line impedance. Formation

of each of the two lines by a pair of coaxial cables adds

shielding and permits a further reduction in impedance

loading by driving the shields as guards at the center conductor potential. The use of two contacts per line not only isolates cells which are not being tested, but, by

proper arrangement of the switches, permits cancellation

of most of the thermoelectric voltage generated in switching.

4) There were several miscellaneous performance re- quirements:

a) uncompensated thermoelectric voltages must be less than 3 nV;

b) leakage resistance must be greater than 10 GU;

c) expected life must be greater than 10 years; and

d) cell circuits must open during power failure. An evaluation of the available switching methods and

hardware showed that many switch designs might be

adapted to meet a majority of the requirements. However,

the multiple-level individually actuated crossbar switch most nearly met the total requirement. The crossbar switch

is advantageous in consideration of the actuating power

level and offers a life expectancy of 10' actuations, which

might approach 100 years in this application. Finally.

the crossbar switch is amenable to modification to over- come its major fault which is the level of possible thermoelectric voltage.

III. CONSTRUCTION

OF THE PROTOTYPE SWITCHING

To demonstrate the feasibility of the selected crossbar

switch, a prototype system has been constructed consisting

of three modules each capable of servicing 10 cells. One

of the modules is expanded to permit use of a local semi- automatic cell selection. These modules employ commercially available crossbar switches that are modified both

in the switch structure and in the materials used in the conducting circuit. Fig. 1 is a simplified diagram of the

6-level switching used for each cell. Levels 1 and 3 are used

to guard the voltage switched on level 2; similarly, levels 4 and 6 are used to guard the voltage switched on level 5. Cell leads are connected to levels 2 and 5, and the guards

are also switched to minimize possible interaction in the system and unwanted noise pickup. Addition of electrically actuated sections to the switch may be employed to expand

the switching for a local or special function. There is also a simple local switch that connects the cell lines to output

terminals and disables the automatic switching function. In any system intended for use in scanning standard cell voltages with high precision, consideration of thermoelectric voltage effects is mandatory and is often the

limiting criterion on the level of precision achieved.

Briefly, the ideal thermoelectrically quiet system would

be entirely of copper. Only slightly less elegant would be

a system employing copper, silver, and gold, which can produce thermoelectric effects of 100 nV/K and for which

a passive thermal shield is normally adequate. Such

materials are not available on crossbar switches now made.

Thus modification of the switch structure and careful design of the module are necessary to achieve the design

requirements.

The crossbar switches employed are modified to use only beryllium-copper conductors and gold-silver-plati- num contacts which are thermoelectrically matched to each other with less than 250 nV/K. This modification

limits the critical junctions to the two necessary connections to copper lead wires in each circuit. In addition to connecting pairs of switches in a bucking configuration

and using cadmium-tin (low thermoelectric voltage)

solder, two methods are employed to minimize thermoelectric effects. These may be most readily understood by

reference to Fig. 2, which shows the internal switch construction, and Fig. 3, which shows the switch actuators. The first method employs a heavier than usual lagging

of the switch environment against laboratory temperature.

Visible behind the 10-cell switch module in Fig. 2 is the containing case formed of two nested aluminum boxes

separated by a layer of foam insulation. The thermal masses of these boxes and of the switch mounting piate.

and the iow conductivity of the foam insulation, are used

to reduce thermal gradients that might be troublesome. 252
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1974

LIVE. 6

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Fig. 1. Simplified switching diagram for each cell.

Fig. 2. Internal switch construction.

Fig. 3. Switch actuators.

Aiding the filtering action is the high thermal conductivity

of the aluminum.

Additionally, critical junctions are heat-sinked inside

close-fitting copper sleeves so that the effects of thermal

transients are minimized. The added thermal mass also

helps to equalize temperatures in the switches. These

sleeves can be seen in Fig. 2, as can the rotary switch used

to set the local system function of the module.

The final method used is to remove nearby heat sources from the vicinity of the circuitry and to shield them by

interposed thermal reflectors. Fig. 3 shows the actuators, which are made remote from the switch proper to minimize

heat input from this source. The switch mounting plate

and the actuator base plate serve as heat reflectors. Heat

from apparatus outside the switch is limited by the

lagging and by the physical spacing required by the case. In the particular module shown in Fig. 3, an extra actuator

set is used to provide a local/automatic-selection mode.

Fig. 4 shows an assembled switch module. A warning

signal is generated when an attempt is made to call a cell and the module has been placed in the local mode. This

signal is used to illuminate an indicator light in this particular module. In the local mode, however, an exclusive lockout function is performed and cells cannot be addressed by the system. Shown also are the rows of coaxial connectors used to input the cell voltages to the module,

to output these voltages in the local mode, and to inter- connect the module to the system. Fig. 5 shows the coaxial connector parts. The core is of copper to minimize thermoelectric voltages, insulation is of Teflon s and noncritical parts are of more common materials. The connector is designed to permit direct connection to a coaxial cable and to preserve the guarding feature.

IV. PROTOTYPE SYSTEM

Assembly of the switch modules into a prototype system

requires addition of both system controllers and measuring

apparatus. To permit operation of the switching, a manually operated system controller and a local controller were constructed. An exclusive lockout is provided so that

5 Registered trademark of E. I. duPont de Nemours & Co. 253
BRAUDAWAY AND KLEIMANN '. HIGH-RESOLUTION PROTOTYPE SYSTEM

Fig. 4. Expanded 10-cell switch module.

n, n4% mm 1 O «

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Fig. 5. Guarded coaxial connectors.

when any cell in the system is addressed on a line, no

operation. Such a limited life is unacceptable. Thus a

additional cell may be addressed on this line until an

intentional release has been accomplished. In the manual

controllers this function is accomplished by sets of inter- locking pushbuttons, but in the computer-controlled

operation the exclusive interlock will be generated in the

interface. Logic employed in operation of the prototype system is of the "address-then-wait-and-see" variety. After a cell is addressed, a callback from the switch indicates completion

of the switching task. In the manual controller, this call- back is used to illuminate an appropriate light. Release

of the cell is also indicated as the light extinguishes.

Polarity of cells addressed on either line is set by a master

polarity switch with a slow-action center-off pattern. All necessary system power is supplied by the controllers. Except for its simpler switching task, the local-automatic

controller is identical in operation to the system controller. After assembly of the total automated system, it is antici- pated that these controllers will be invaluable in diagnostic

testing of individual modules. The design of a system for high-precision cell voltage measurement entails many considerations, and effort on

this phase of the project has been addressed only toward

solving some of the more critical problems. One major

problem is the switching; which, by the nature of the measurement process, is actuated much more often than

that associated with scanning. For example, as might

occur in a null-balancing mode of operation, a switch actuated at a rate of 1 per second might exhaust an

expected lifetime of 107 actuations in only 100 days of measuring logic was chosen in which a null-balance is not

obtained and switching is done only to provide 10-txV increments of the voltage difference between the two

cells under test. The remaining difference voltage is amplified and digitized directly using a solid-state conversion with an essentially unlimited life expectancy. Such

an approach, which is similar to the older deflection methods, requires that the amplifier have a high input

impedance to minimize cell loading to a negligible level and provide for accurate measurement. To complete the prototype of the automated standard

cell measuring system, a rudimentary measuring facility was assembled using a precision potentiometer to supply

the 10-juV increments. A commercial precision amplifier

with digitized output was obtained and modified so that

the resolution approaches 10 nV. The input impedance of

this amplifier transiently starts at 8 kQ, and rises rapidly

to significantly greater than 10 M£2. This impedance level

is sufficient to offer negligible cell loading, lower in fact than can be achieved with a normal light-beam galvanometer and a careful approach to near null-balance.

V. RESULTS

At the time of writing, over a year's experience has been

gained in daily use of the prototype standard cell test

facility described. Operation has been in the manual mode

with visual readout from the digital display because the

computer-controller facility is not yet available. During

this period, problems have been due to 1) unfamiliarity

with the equipment and with provided operational modes,

2) adjustment problems with the crossbar switches as- sociated with the modified structure, and 3) difficulties with ancilliary equipment used to complement the rudi- mentary measuring system and with cabling interconnections. All of these problems are in the category of expected

difficulties, and none has been major.

Estimates of the switching capability based on the

testing of component parts indicated that a residual un- compensated thermoelectric voltage level of 1 nV/K had

been achieved. Extrapolating to use in a laboratory en- vironment with uncontrolled but modest nearby heat

sources indicated that the 3-nV performance level of the

design goal had been met. Shown in Fig. 6 is a bar-graph

plot of the results of an offset voltage measurement on all active positions of the prototype system in the use condition. Contributions to the plotted figures come from the

switching, the interconnecting cabling, and, not insignifi- cantly, the equipment used to perform the special test. It is significant that the mean assignable offset to these measurements is only about 3 nV. Data obtained using the prototype system indicate that the overall error in measurement that can be assigned to the system proper is less than 10 nV, within the design

goal. Also demonstrated in operation of the system is the

flexibility of the scanning in fulfilling its intended function. Thus a system can be built which displays measure254
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-23, NO. 4, DECEMBER 1974

8

8 "

9 i 2 3 k 5 6 T

Fig. 6. Magnitude of measured offset voltages in prototype system.

ment capability at the limit of that now envisioned, which

is readily flexible, and which is adaptable to new approaches. It should be pointed out that not all of the

provisions of the system to ensure measurement excellence have been employed. Specifically, the guarding feature has not been activated and has been employed only as a

shield. As the system expands and the workload is in- creased, however, such a feature can be used to ensure

that highest measurement accuracy is maintained.

REFERENCES

[11 G. W. Vinal, Primary Batteries. New York: Wiley, 1950, ch. 6.

[2J G. D. Vincent, "Construction and characteristics of standard

cells," IRE Trans. Instrum., vol. 1-7, pp. 221-234, Dec. 1958.

[3] W. G. Eicke, Jr., Nat. Bur. Stand., Washington, D. C., private

communication, unpublished computer programs for reducing standard cell data.

[4] B. F. Field, T. J. Finnegan, and J. Toots, "Volt maintenance at NBS via 2e/h: A new definition of the NBS volt," Metrologia,

vol. 9, no. 4, pp. 155-166, 1973.

[5] D. W. Braudaway, An Automatic Standard Cell Comparator, Sandia Labs. Tech. Manual SCTM 129-62(24), 1962.

[6] , "Automatic testing of standard cells," ISA Trans., vol. 3, no. 5, pp. 211-216, 1966.

[7] H. Hirayama and Y. Murayama, "Automatic measuring system

for a control of standard cells," IEEE Trans. Instrum. Meas.,

vol. IM-21, no. 4, pp. 379-384, 1972.

[8] W. G. Eicke and J. M. Cameron, "Designs for surveillance of the volt maintained bv a small group of saturated standard

cells," Nat. Bur. Stand." Tech. Note 430, 1967.

255
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-23, NO. 4, DECEMBER 1974

Standard Cell Enclosure with 20-uK Stability

ROBERT D. CUTKOSKY, fellow, ieee, and BRUCE F. FIELD, member, ieee

Abstract—Groups of standard cells are needed to maintain the

National Bureau of Standards (NBS) volt between periodic re- assignments via 2e/h. An enclosure to hold six standard cells has been designed and three enclosures have been assembled. The

enclosures consist of four concentric aluminum cylinders separated

by foamed polystyrene insulation. Two of the four cylinders are controlled, an outer one by a platinum resistance thermometer and dc

amplifier and the inner one at 30°C by a second platinum resistance thermometer and an ac transformer-ratio bridge followed by a phasesensitive detector, a dc amplifier, and a heater. The innermost of the four cylinders is a thermally lagged cell compartment. The

temperature excursions over periods of at least several days were

found to be less than 20 mK, thus having a negligible effect on the standard cell EMF's.

I. INTRODUCTION

THE NEED FOR a stable and reproducible standard

of EMF led to a redefinition of the National Bureau of Standards (NBS) volt in terms of 2e/h via the ac Joseph- son effect. This decision eliminated the dependence of the NBS volt upon the long-term stability of a large group of saturated standard cells. However, the 2e/h measurements

are relatively difficult and time consuming; thus it is impractical to make measurements frequently (e.g., every

day). Groups of standard cells are therefore needed to maintain the volt between periodic 2e/h measurements. The EMF's of these group must be predictable to within a few parts in 108 for up to several months. For saturated standard cells, such predictability requires that the cell temperature be stable or predictable to about 200 mK and

that temperature differences between cell electrodes be

constant within 30 nK. Accurately measuring the cell temperature is difficult, because the cell temperature lags the

enclosure temperature by some unknown amount. Thus to introduce negligible error due to temperature variations, the total temperature fluctuations should be less than

200 mK- An enclosure to hold six standard cells in air was

designed, and three enclosures have been assembled. A

double oven system is used with separate proportional

controllers for each oven. The mechanical characteristics

of the enclosures are described, the controller circuits discussed, and data on the performance of the systems are presented.

II. CONSTRUCTION DETAILS

The enclosures consist of four concentric aluminum

cylinders separated by foamed polystyrene insulation

(Fig. 1). The outer cylinder thermally lags the enclosure from ambient temperature changes and reduces temperaManuscript received July 3, 1974; revised August 20, 1974. The authors are with the National Bureau of Standards, Gaithersburg, Md.

CONTROLLER LEADS 1 Fig. 1. Side view of 32-cm-diameter enclosure, cut away to show

interior cylinders.

ture gradients over the surfaces of the enclosure. The outer oven is heated by resistance wire helically wound on the

outside of the cylinder. No heating is applied to either the top or bottom surfaces because the thickness of the alumi- num serves to distribute the heat evenly and to minimize

temperature gradients. The temperature of the oven is sensed using a miniature platinum resistance thermometer

imbedded in the aluminum wall. This oven has two chambers, one in which the inner oven and cells are contained,

and another in which temperature-sensitive circuit components and a cell EMF reversing switch are mounted. The

inner oven also has a heater that is wound on the cylinder

wall only, with no heating applied top or bottom.

The cell compartment is thermally (and mechanically)

connected to the inner oven at one point. At this point the

cell EMF leads enter the compartment and are thermally

tied to the cylinder. Temperature is also sensed at this interface between the two enclosures by a miniature plati- num resistance thermometer. The cell compartment and

the inner oven are separated by the polystyrene insulation everywhere, except for the one location previously mentioned; therefore, almost all heat flow will be through this interface between the cylinders. By sensing the temperatures at this interface and applying the heat to the inner oven, a condition of zero (negligible) heat flow into the unheated cell compartment is obtained. This condition

assures that temperature gradients in the cell compartment

are negligible. An auxiliary 25-Q platinum resistance thermometer is mounted in the cell compartment to permit

monitoring the temperature with an external bridge.

Reduction of heat leaks into the cell compartment was

of primary inportance. Small-diameter wire (30 gauge)

256
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1974

and a series of special thermal tie-downs were used to shunt

heat flow along the incoming leads to the more massive

aluminum cylinders. Standard cells are noticeably dis- charged by a resistance across the cell even as large as 109 12; therefore only tie-downs with very high electrical resistance can be used. The tie-downs consist of a boron

nitride washer and a copper washer held firmly to the aluminum surface by a Nylon screw. The wire was wrapped

under the head of the screw and then passed through a hole

in the screw to the other side of the aluminum surface. This arrangement provides low thermal resistance but

high electrical resistance (1012 Q) between the wire and the aluminum. Both cell EMF leads and temperature control

leads were tied down at several places in the enclosure.

III. CIRCUIT DETAILS

The electronics used to control the temperatures of the two cylinders are housed in a separate package with pilot

lamps to indicate the status of the enclosure. There are three controllers, one for the outer oven, one for the inner

oven, and a completely independent backup controller for the inner oven. The two main controllers for the ovens

are bridges with high gain amplifiers to provide proportional control. Power to the main controllers is normally

supplied by a well regulated commerical power supply.

Line voltage variations are essentially eliminated and have no effect on the controllers. The backup controller has its own power supply and a mercury thermostat for on-off type control. Loss of temperature control for only a few hours can

cause EMF changes in standard cells for up to several weeks. Control must be maintained even during ac power

failures. Mercury batteries have been included that will power the enclosure for two days (in a 23°C ambient)

with less than 100 n~K change in the cell temperature.

Switchover to battery power occurs within milliseconds of a power failure, with large valued capacitors supplying

power during the switching time. Transfer back to ac power occurs 10 s after restoration of power to allow power

line conditions to stabilize. The controller for the outer oven consists of a 500-fi platinum sensor in a Wheatstone bridge and a dc amplifier driving a 60-12 heater. The dc amplifier is a single lowvoltage-offset operational amplifier with a high power

output stage. Finite gain of the amplifier causes a temperature change in the oven of 0.002°C for a 1°C change in ambient temperature. Operation is at 29.7°C with 4 W of power required for an ambient temperature of '23°C. Since

the amplifier is capable of supplying 15 W to the enclosure,

it can maintain temperature control down to — 20°C. The inner oven, with typically 0.1 W applied, is held at a temperature of 30°C, 0.3°C higher than the surrounding

outer oven. Requirements for the controller are long-term

stability, high sensitivity, low noise, and low sensitivity to ambient temperature changes. A platinum resistance thermometer was chosen as the sensor because of its excellent long-term stability. Power dissipation in the sensor imposes a practical limitation on the maximum signal

obtainable from the sensor. As sensor dissipation is in- creased, the sensor temperature becomes higher than the temperature of the controlled surface. This temperature

difference must be maintained constant for best stability of the controlled surface. Thus larger signal output requires

higher sensor dissipation, which in turn requires better amplitude stability of the bridge excitation voltage. A

compromise of a 100-12 (at 0°C) thermometer with 1 MA

of thermometer current was chosen. Although several high quality operational amplifiers

have very good dc characteristics (low voltage offset, current offset, etc.), these were judged inadequate for this system. It was decided to use ac excitation on the

bridge, ac amplifiers, and a phase-sensitive detector. Using

ac excitation permits the use of a transformer-ratio bridge

instead of a classical Wheatstone bridge. Advantages of

this arrangement are that the transformer-ratio arms have

a negligibly small temperature coefficient and so do not have to be thermostated. The inclusion of only the refer- ence resistor in the outer oven produces a circuit that is negligibly affected by changes in ambient temperature. A

two-stage transformer is used for a four-terminal measure- ment of the thermometer resistance so that changes in the

lead resistance have no effect. The addition of an auxiliary transformer allows for easily adjusting the temperature

(a feature used in setting up the enclosure). A 100-12 standard reference resistor is used with the 110-12 platinum

thermometer (at 30°C).

As noted before, the bridge excitation voltage must

remain constant. The circuit diagram of the oscillator used

is shown in Fig. 2. A Wein bridge oscillator produces a sine wave of approximately 400 Hz. A dc signal proportional to the average value of the output voltage is generated, and

this signal is then compared to the voltage of a temperature-compensated reference Zener diode. The resulting error signal is fed back to the oscillator to stabilize the out- put voltage at 2.8 V. Observed changes (drifts) of output were less than 0.005 percent with negligible effect from

temperature changes. An additional amplifier was added

to the circuit to provide two outputs 180 degrees out of phase with each other, because both are needed for the

phase-sensitive detector. The bridge excitation transiormer Tl is a conventional

two-stage device with a toroidal Mumetal shield between

stages. Auxiliary transformer T2 is coupled to it and pro- vides a means of adjusting the effective transformer ratio, which determines the control temperature. The details of these transformers and the remaining circuitry are shown in Fig. 3. The low level signal from the bridge is applied to a transformer (T3) with a tuned output. This transformer matches the low impedance of the bridge (» 50 12) to the optimum sources impedance of the first amplifier (107 0). The transformer output is tuned to 400 Hz to reduce 60-Hz

noise pickup. The two operational amplifiers that follow have a total gain of 90 dB. This amplified signal is fed to a

phase-sensitive detector, then to another operational

amplifier. The dc output of this last stage drives the 300042

257
CUTKOSKY AND FI KLD '. STANDARD CKLL KNCLOSURK

PSD

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1001. but

Fig. 2. Circuit diagram of oscillator used to drive transformer ratio bridge for inner oven.

Fig. 3. Complete circuit diagram (except for oscillator) of inner oven controller. (Numbers on transformers indicate number of turns.)

heater on the inner oven, and the heater voltage is in- dicated on a meter mounted on the front of the electronics package. Since the heater voltage is proportional to the oven temperature, the meter can be calibrated in degrees. For this system the total range of the meter is approximately 1000 mK; thus 10 fiK can easily be resolved. Buffering of ambient temperature by the outer oven

ensures that the inner oven operates at almost constant power. Therefore a substantial change of the inner heater

voltage can be interpreted as a failure of either or both

controllers. The meter that indicates inner heater voltage

is a meter relay that can be set to close a switch contact if the voltage changes (equivalent to a temperature change

of 300 fxK) . If this happens, 110 V ac power is removed

from both main controllers and applied to the backup controller. This controller has a mercury thermostat in contact with the inner oven. The thermostat opens and closes a

sensitive relay that applies half-wave rectified 1 10-V ac to the 3000-ft inner heater. Power for the mercury thermostat

is from a dc supply independent of the main supply for the

other controllers. The backup controller will maintain the temperature to within 0.0 1°C of nominal until manually

reset.

IV. ADJUSTMENT

Several modes of operation are possible to allow initial adjustments of the controllers. First the mercury thermostat controller is used and the operating temperature measured using an external bridge and the platinum ther- mometer in the cell compartment. The mercury thermostat

controller is then disabled and both main controllers are turned on. The temperature of the outer oven is adjusted so that the inner heater operates at 0.1 W. This adjustment

is accomplished by shunting one of the arms of the bridge.

Trimmer capacitors are added to the two ac amplifiers to make the total phase shift in the amplifiers zero. The

trimmer capacitor on the tuned transformer is adjusted for a maximum 400-Hz signal. Additionally, a small trimmer

capacitor is shunted across the reference resistance to balance the capacitance in the platinum thermometer. The

temperature of the inner oven is adjusted to be within

0.0 1°C of the mercury thermostat temperature by picking proper taps in the ratio transformer and, if necessary, by

shunting the reference resistor with a large valued trim

resistor. Once the temperature (heater voltage) stabilizes, the system is reset so that the backup controller will operate if required.

V. PERFORMANCE

The performance of the enclosures was evaluated by

monitoring the temperature changes in the enclosure with

time, and changes due to ambient temperature changes.

An ac resistance thermometer bridge fjl] was used to compare the resistance of the 25-f2 thermometer in the cell compartment to a low-temperature-coefficient thermally

lagged resistor. A plot of the cell temperature versus time for one of the

enclosures is shown in Fig. 4. During this test the enclosure was in a laboratory environment where ambient temperature fluctuations were less than ±1°C. A check of the temperature two months later showed no change within

experimental error (^50 nK, due to drift of the reference

resistor). Temperature drifts this small produce EMF

changes in standard cells that are negligible when compared to typical drifts of the EMF's of cells. To determine the sensitivity of the enclosure to ambient

temperature, both the enclosure and electronics package

were placed in a temperature-controlled chamber. Temperature was monitored by the ac thermometer bridge

which was kept at 23°C. The data obtained are plotted in

Fig. 5. The test was started with measurements at 23°C. The chamber temperature was then raised to 29°C, and

after allowing the enclosure to stabilize for one day, measurements were made on three different days. The

chamber temperature was then lowered to 16°C and

another measurement was made after a one-day wait. Similar measurements were made for 19°C and again for 23°C. The test took a total of 10 days and the repeat measurements at 23°C do not indicate any significant drift during the test. The data are reasonably linear; thus the

effect of ambient temperature on the enclosure is the slope

of the least squares fitted line, or —4 X 10~fi

. A test for sensitivity to temperature gradients was per- formed by placing a 40-W light bulb in contact with the upper edge of the enclosure. No detectable change was observed during the 8-h monitoring period in cither the tem258
IKK K TRANSACTIONS ON I NSTRUM KNTATION AND MKASURKMKN'T, DKCKMBKR 1974

Fig. 4. Deviations of cell temperature from nominal as monitored by independent thermometer bridge.

62 8

62 7

62 6

"i 1 1 T

I 0.1 pV

SO = 0 017 uV

J L

2 3 MONTHS

Fig. 6. Mean EMF of six cells in one enclosure minus mean EMF

of six cells in second enclosure.

1 1 1 1 1 1 1 1

a£3 lOO -

19| *50 -

2 •

^ -50 -

2

- ,0 ° -

I 1 I I I 1 I I 16 16 20 22 24 26 28 30 AMBIENT TEMPERATURE CO

Fig. 5. Temperature change in cell compartment as a function of ambient temperature cf enclosure and electronics package.

perature of the auxiliary thermometer within the cell compartment or in the temperature of an additional plati- num thermometer temporarily installed near the top of the cell compartment.

To complete the evaluation, standard cells were mounted in the enclosures and regularly compared. Fig. 6 shows the mean EMF of six cells in one enclosure minus the mean EMF of six cells in a second enclosure. The standard

deviation of the least squares fitted line is 0.017 uV. The

measurements were made potentiometrically with re- dundant measurements made to evaluate the random

error. The typical standard deviation for the difference

between the mean EMF's was 0.003 ^V. Additional dayto-day random errors are estimated to be 0.007 uV. The

root-sum-square total of 0.008 uV is only about one-half the standard deviation observed. Additionally, individual

cells occasionally showed unexpected changes as large as 0.06 uV. Temperature fluctuations in the enclosure are too

small to account for this, so it is spectulated that residual chemical processes in the standard cells are responsible for the additional "noise."

VF CONCLUSIONS

It has been demonstrated that an air enclosure can pro- vide outstanding temperature control for a small group of standard cells. The use of a double oven enclosure, ac

excitation, and a transformer-ratio bridge eliminate the problems due to ambient temperature changes. A platinum

resistance thermometer (while having less sensitivity than some other sensors) is still suitable where long-term temperature stability is desired. The major limitation on the

stability of standard cell EMF's has now been shifted from

temperature measurement errors to EMF measurement

errors and the inherent instabilities in the cell. REFERENCES

[1] R. D. Cutkosky, "An a-c resistance thermometer bridge," J.

Res. Nat. Bur. Stand., vol. 74C, pp. 15-18, Jan. 1970.

Reprinted by permission from IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT

Vol. IM-23, No. 4, December 1974, pp. 295-298 Copyright 1975 by the Institute of Electrical and Electronics Engineers, Inc. PRINTED IN THE U.S.A.

259
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-23, NO. 4, DECEMBER 1974

Standard Cell Cali bration via Current Transfer

EDWIN R. WILLIAMS, PAUL T. OLSEN, and BRUCE F. FIELD, member, ieee

Abstract—The EMF's of standard cells are now being transferred between laboratories over a cable with a precision of 4 parts

in 10s to provide an instantaneous comparison of the 2e/h and

7P

' experiments being carried out at the two facilities. This is accomplished by transferring a constant current that produces a 1-V drop

across standard resistors located at both ends of the cable.

I. INTRODUCTION

IN THE Electricity Division of the National Bureau of Standards (NBS), there are three experiments in progress that can be used to monitor the volt, the ohm,

and the ampere. The ac Josephson effect has proven ex- tremely useful in monitoring the unit of voltage in terms

of the fundamental constant 2e/h [1]. The calculable capacitor is now providing an absolute determination of the ohm (Onbs/A) [2]. And the gyromagnetic ratio of the proton (7/) can be used to monitor the unit of cur- rent [3]. In addition to providing this essential monitoring

service, these three experiments can be combined with

a few other fundamental constants to provide an improved

value of the fine structure constant a. The following equation shows this relationship [4]:

a-2 = (l/4ie.)(c£2/aNB s)(MP'/MB)C(2e//l) NB S/(7p')NBs]

where Rx is the Rydberg constant for infinite mass, c is the speed of light in vacuum, and mp'/mb is the magnetic

moment of the proton in Bohr magnetons. The subscript

NBS implies that (2e//i)NBs is measured in terms of the NBS unit of voltage and that (7p')nbs is measured in terms of the NBS unit of current. Because these three experiments are being done in a coordinated effort, we can

reduce the uncertainties normally introduced in transferring the electrical units. However, the yp experiment

is being performed in the nonmagnetic facility which is located H km away from the 2e/h and absolute-ohm

laboratories. In this paper we describe our efforts to reduce

the uncertainty of the voltage transfer so that it negligibly

affects the y p experiment. We also suggest how these techniques may be applicable to other problems in the

precision electrical standards field. In the 7P

' experiment the precession frequency of pro- tons in a sample of water (cjp

') is measured via a nuclear magnetic resonance (NMR) experiment, in which ui p =

yp'B, where the prime denotes that a spherical water sample is used. A known magnetic field B is produced by passManuscript received July 3, 1974; revised August 8, 1974. This work is a contribution of the National Bureau of Standards, and is not subject to copyright.

The authors are with the Electricity Division, National Bureau

of Standards, Washington, D. C. 20234.

ing a 1-A current through a solenoid whose dimensions

have been carefully measured. In practice, the 1-A current

is produced by making the voltage drop across a precision

1-0 resistor equal to the EMF of a standard cell. For this experiment the EMF of the cell and resistance of the resistor need to be known in NBS units to determine yp

'

in NBS units. Two different experimental arrangements

were used to transfer the cell EMF for comparison with

the NBS volt. The first was a 10-mA system with 100-O

resistors at each end used to transfer the voltage between

two standard cells. The second uses a 1-A system to transfer the unit of current directly. This system can also provide a simultaneous voltage transfer. The first system is the one most likely to be useful to others because it is easier to build. We are using the second method because

in the yp experiment it is the unit of current that is re- quired. We will describe the present 1-A arrangement

first, then describe the 10-mA system because it has a

guarding system that should be of general interest.

II. PRESENT METHOD OF TRANSFER

In our present experimental arrangement, we directly

transfer current by placing the solenoid in series with the

long cable and with the 1-0 standard resistor Ri which is located in the 2e/h laboratory. (See Fig. 1.) This arrange- ment does not noticeably affect the quality of the NMR

experiment. We need only make sure that the leakage

current is sufficiently small. In addition we have a second

142 resistor R2 in the yp laboratory which allows us to do a

voltage transfer at the same time that we are doing the NMR experiment.

The experiment is performed by one operator adjusting

the constant current source so that the EMF across R2 is equal to Eyp -. A second operator adjusts and reads the frequency of the NMR system. These two operations are being carried out in the same laboratory so that when

both are balanced we communicate by intercom to the

2e/h laboratory where the EMF across resistor Ri is measured. Then the current is reversed in the entire system, and the triple balance procedure repeated. This

reversal has two purposes. First, any residual earth's

field is averaged out in the NMR data and second, any

thermal EMF's in the 1-0 resistors are eUminated. In the NMR part of the experiment, a frequency o) p

is measured

which is proportional to the current 7. The ratio Ei/wp

is independent of the current I and is the quantity of

interest in the yp experiment [_y p = (u3p'/E\) (Ri/Kc )

,

where Kc is the coil constant]. We, therefore, do the

transfer using slightly different currents, i.e., different values of E7r and the statistical variation in the ratio

260
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1974

SOLENOID

CONSTANT

CURRENT Fig. 1. Current transport system that unites 2e/h and y p laboratories. Resistors R\ and Ri are in series with the solenoid. With 1 A flowing, up' can be measured simultaneously with E\.

Ei/

u

P

' will give an unbiased measure of the precision of the transfer. The standard deviation of a set. of measure- ments is usually about 0.04 ppm. Fig. 2 shows a composite

of recent Ei/up

' measurements. The day-to-day variations are larger than would be expected from the uncertainty

of each measurement. We feel this is not caused by in- adequate transfers, but by systematics in the NMR

experiment. Namely, there is presently about a 1 part

in 107 uncertainty in the way in which we define the center frequency of the NMR signal. This must be done at the

start of each day's set of measurements. We expect to re- duce this cause of uncertainty in the near future. The

straight line shown in Fig. 2 is the drift rate of E\/wP

' pre- dicted from early measurements of the coil constant Kc [3]. It is still too early to say if the predicted drift rate is in agreement with this NMR data. The voltage transfer is also accomplished during the measurement process. We do not need this transfer value

for the yp experiment, but it is helpful as a short term

standard. For example, if the 2e/h laboratory is busy one

day, we can still do a yp experiment using these cali- brated cells. Or when testing systematic effects in the yp experiments, we need not bother using the 2e/li cells. The value of the yp cells is given by

Eyp

i = EdRt/Ri).

The ratio Ri/R\ was measured by the resistance cali- bration section. We also measured it by interchanging R\

and Ri and doing another transfer. The ratio we obtained

for Ri/Rx disagreed with the calibrated values by only

0.02 ppm which is within the uncertainty of the measurements. This agreement gives us further confidence that the transfer system is working properly. Fig. 3 shows the EMF of the average of four cells located in the yp lab over

I —l 1 i 0 10 20 SO 40 TIMElDaysl Fig. 2. Ei/u,,' versus time. The measured quantity £i/u,' is in- versely proportional to The error bars represent the 1-standard-deviation uncertainty of each day's transfer. The straight line

is the drift rate of —0.0157 ppm/day predicted from dimensional measurements of the solenoid.

-48

i i r

1

y', STAN0AR0 CELLS vs. TIME

1 ,- T

1

• o

o

CO • •

-47•

Ippm

CEL1 -46

1970 , 1971 , 1972

, 1973 , 1974 0 10 20

1 I 1 TIME (Yearsl (Oaysl

Fig. 3. Average of four standard cells maintained at y p

' laboratory

versus time. The transfers through 1973 were made by direct cell transport taking 3 weeks per point. The June 1974 value is a composite of six transfers taking 30 min a piece.

the past four years. The last point in June 1974 is an aver- age of the six transfers performed by the new technique.

The earlier transfers were conventional cell transports

which took about 3 weeks to obtain each point, and the

results were not always satisfying. In our present scheme

each transfer takes 30 min.

III. LEAKAGE CONSIDERATIONS

AND GUARDING

In the present configuration (Fig. 1), leakage from the

center conductor to the return shield of the two cables is the leakage which affects the yp measurement. There is an average of 11 V across the long cable with 1 A flowing

and a leakage resistance of 2 X 1010 0. In the 100-m cable

there is a 40-V drop because of the solenoid resistance. Again this leakage resistance is about 2 X 10'° Q. Therefore the correction for leakage current is only about 2 parts in 109

. In the voltage transfer any leakage from the

shield of the coax to ground causes an error. We have a

special cable which has one center conductor and three

coaxial shields, each electrically insulated from the other. The innermost shield is used to carry the 1-A current be- cause the resistance from it to the next shield is 4 X 109 Q. The average 4-V drop across this cable produces only a small correction, due to leakage, to the volt transfer. If the cable leakage degrades in time or if higher accuracies

261
Williams et al.: standard cell calibration

become necessary, a guarded system can be employed to further reduce the leakage in this 1-A system. A guard

system was employed in our earlier 10-mA system.

IV. THE 10-mA TRANSFER SYSTEM

This transfer (see Fig. 4) is performed by sending a constant current (10 mA) through two 100-ft resistors located

at either end of the cable. The 1-V drop across each re- sistor is simultaneously compared with two sets of standard

cells located near the two resistors. This simple procedure can be completed with a precision of about 2 in 108 in only a few minutes. In this 10-mA system the outputs

of both galvanometers are simultaneously recorded, and

then the recordings are compared for computation of the double balance condition. This procedure reduces the problem of drift in the galvanometer. In addition, the noise and drift are correlated in both recordings, and with proper

analysis, the limitations produced by the noise of the

current source can be reduced. For this reason we obtained better precision in the 10-mA system than in the 1-A system.

To prevent leakage of current in the long transfer

cable, a dynamic guarded system is used (Fig. 4). The

transfer current is sent down the center conductor. The

first coaxial shield has a guard current in it such that the

potential difference between the guard and the center conductor is less than 0.01 V along the entire length of the cable. This reduces the leakage to less than 2 parts in 10 10 of the total current. Both currents are returned on a second coaxial shield. A third coaxial shield is grounded

to help reduce pickup problems. In this 10-mA setup the accuracy of the transfer is limited to the accuracy with which we measure and maintain the ratio of the two

100-ft resistors, or about 5 parts in 108. Two inductors (0.3 H each) not shown in the drawing were included at either end of the cable. The capacitance

of the cable combined with capacitors placed across the 10042 resistors formed a t filter which reduced the noise

across the 100-Q resistors. One must choose inductors

that are not saturated by the 10-mA current. In the 1-A

system no additional chokes seem necessary.

V. ADVANTAGES OF CURRENT TRANSFER

A current transfer system is preferable to a direct voltage transfer when long cables are involved for the

following reasons.

1) The thermal EMF's along the cable have a negligible effect on the transfer; i.e., the thermal EMF's

need not be evaluated.

2) The lower impedance of the system reduces the

cable leakage problem and reduces the amount of elec- tronic "pickup" noise.

3) A choke can be effective in further reducing any

remaining electrical interferance noise before it is "seen" by the standard cell. With our 1-A system the major advantage is that we

Fig. 4. Diagram of cell comparison by current transport that in- cludes a "dynamic" guard system.

need not maintain a high quality voltage standard at the 7P

' laboratory. A large effort is involved in the maintenance of the cells in the 2e/h laboratory, for they form

the basis of the U.S. legal volt. It is logical to take advantage of this effort by directly using these cells in measuring

the current used for the yp

' experiment. One final ad- vantage is that the EMF developed across the transfer

resistors is a very rugged source. Unlike standard cells,

it can be shorted without damage or the voltage can be

looked at on an oscilloscope without using a very high

impedance probe.

VI. POSSIBLE USES OF THE

CURRENT TRANSFER

This transport technique could readily be adapted to a dissemination system which provides a network of substations. Each substation would have continuous cali- bration via current transfer back to the master reference. Such a system would be useful in standards laboratories or in industrial applications where precise standards are required in a production line environment. Each station could provide any voltage, for example, from 0 to 10 V. The main system could easily be safeguarded against in- experienced operators, and stations having accuracies ranging from 10 ppm to 0.03 ppm could all be on the same

system.

A prototype station which provides this type of ser- vice is now being maintained in the Electricity Division

of NBS at the 10-ppm level of accuracy. It is a self-service unit with a range of 0 to 10 V and is being used for digital

voltmeter calibrations by the NBS staff. Additional uses for our transfer cable at the nonmagnetic

facility will be to provide a transfer for the absolute-volt experiment presently in progress at the facility. An abso- lute-ampere experiment is in the planning stages and this cable will also be used to transfer this measurement to the 2e/h laboratory.

VII. CONCLUSIONS

Current transport has increased the accuracy with which we can determine yp

' in the NBS units and at the same

time has reduced the burden of maintaining a high quality

voltage standard at the nonmagnetic facility. We can now measure the current in our solenoid with an accuracy on the order of 0.05 ppm in terms of the NBS electrical units as maintained via fundamental constants. This

allows us to draw definitive conclusions about the stability of our solenoid dimensions or NMR measuring system.

262
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-23, SO. 4, DECEMBER 1974

This system demonstrates the feasibility of transferring

precision voltages over even longer cables with accuracies comparable to the present state of the art of maintaining

standard cells in terms of 2e/h. ACKNOWLEDGMENT

The authors would like to thank T. E. Wells for his measurement of the resistances and the power coefficients of the resistors.

REFERENCES

[1] B. F. Field, T. F. Finnegan, and J. Toots, "Volt maintenance at NBS via 2e/h: A new definition of the NBS volt," Metrologia,

vol. 9, p. 155, 1973.

[2] R. D. Cutkosky, "New NBS measurement of the absolute farad and ohm," this issue, pp. 305-309.

[3J P. T. Olsen and E. R. Williams, "A more accurate determination

of yp

' through improved dimensional measurement techniques,"

this issue, pp. 305-309.

|4) B. N. Taylor, W. H. Parker, and D. N. Langenberg, " Determination of e/h using macroscopic quantum phase coherence in super- conductors: Implications for quantum electrodynamics and the fundamental phvsical constants," Rev. Mod. Pkys., vol. 41, p. 491, 1969.

Reprinted by permission from IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT

Vol. IM-23, No. 4, December 1974, pp. 299-302 Copyright 1975 by the Institute of Electrical and Electronics Engineers, Inc. PRINTED IN THE U.S.A.

263
JOURNAL OF RESEARCH of the National Bureau of Standards-A. Physics and Chemistry

Vol. 76A, No. 4, July-August 1972 The EMF-Temperature Coefficient of "Acid" Standard Cells

of the Saturated Cadmium Sulfate Type from 15 to 40° C

Walter J. Hamer, Anna Skapars, and Bruce F. Field

Institute for Basic Standards, National Bureau of Standards, Washington, D.C. 20234

(

(February 9, 1972)

This paper gives the results of new measurements of the effect of temperature on the electromotive

force of standard cells of the saturated cadmium sulfate type. Measurements were made over the temperature range of 15 to 40 °C. Twelve cells of NBS manufacture and twenty-four cells supplied by

two different commercial manufacturers were used in the studies. Final results were analyzed by the method of least squares using computer programs. The relation between emf and temperature for NBS,

company 1, and company 2 cells was found to be given, respectively, by:

E, = E20 °C - [40.44 (t- 20) +0.921 (/-20) 2 - 0.00866 (t-20) 3

] x 10-6 V, E,= £2 o -c- [40.14 (t- 20) +0.888 (t-20) 2 -0.00668 (t-20) 3

] x 10"6 V, Et = E20 - c - [39.28 (t-20) +0.986 (t-20) 2 -0.00943 (t-20) 3

] x 10-« V.

Values for the entropy and heat-capacity changes for the cell reaction in "acid" standard cells are

also given.

Key words: EMF-temperature coefficients of standard cells; standard cells; entropy changes in standard

cells; heat-capacity changes in standard cells; effect of acid on standard cells.

1 . Introduction

In a recent paper [l] 1 the thermodynamics of

standard cells of the saturated cadmium sulfate type were given. That paper dealt with so-called "neutral"

standard cells to which no sulfuric acid is intentionally

added when the cell is constructed. The electromotive

forces (emfs) were known in terms of the legal volt maintained by the National Bureau of Standards and

the emf-temperature coefficient was that derived from

the international formula [2, 3, 4] but converted to the

^69 and <68 scales of voltage and temperature [1], respectively.

It is now common practice to add sulfuric acid in small amounts, between approximately 0.03 and 0.1 /V (or 0.015 and 0.05 M), to standard cells to reduce

the formation of the basic mercurous sulfate,

Hg20 • Hg2S04 which is formed in the hydrolysis of mercurous sulfate according to the equation:

2 Hg2 S04 (s)+ H2 0(1) ?± H&O • Hg2 S04 (s) + H2 S04 (aq) (1)

where s = solid, 1= liquid, and aq= aqueous. Gouy [5] and Hager and Hulett [6] found that the equilibrium

concentration of H2SO4 formed in the hydrolysis of mercurous sulfate (eq (1)) was 0.002 TV while Craig

1 Figures in brackets indicate the literature references at the end of this paper.

and Vinal [7] found it to be 0.00198 TV at 0 °C and

0.00216 N at 28 °C thus showing that it did not change

appreciably with temperature. Cadmium sulfate, the

solute used to prepare the solution in a standard

cell also hydrolyzes to form sulfuric acid, the equilibrium concentration being 0.00092 N at 25 °C [8]. This

concentration is insufficient to prevent hydrolysis

of mercurous sulfate although sometimes cadmium

sulfate, especially if prepared from cadmium nitrate and sulfuric acid or digested with sulfuric acid, contains occluded sulfuric acid in sufficient amount to prevent the hydrolysis of mercurous sulfate and is therefore present through inadvertance. Usually,

however, sulfuric acid in small amount is added to the

cadmium sulfate electrolyte when the cells are constructed. Solutions 0.03 to 0.06 N with respect to sulfuric acid are generally used because mercurous

sulfate exhibits a minimum solubility in the range from

0.04 to 0.08 N. Higher concentrations of acid cause

excessive gassing at the cadmium-amalgam electrode.

Although the effect of the addition of sulfuric acid on the emf of saturated standard cells is well known

[8, 9, 10, 11, 12, 13] published results do not agree as

to the effect of the addition of sulfuric acid on the emftemperature coefficient. Obata and Ishibashi [14]

determined the emf-temperature coefficient of acid

cells over a wide range of acidities, namely, 0.078

to 0.756 N and for temperatures from 15.67 to 29.57

°C and concluded that "the use of the acid electrolyte

264
has very little effect on the temperature coefficient of the cell; the cell containing an electrolyte acidified to 0.76 N [too high for emf stability] has a temperature

coefficient only 11 percent larger than the neutral one." For 0.078 N their value of -42.5 fxY for dE/dT

at 20 °C may be compared with —40.64 /xV for the

neutral cell [1]; it is evident, therefore, that their

results even for low acidity show that addition of acid

increases the emf-temperature coefficient. However, Vosburgh and Eppley [15] stated that a

cell made with saturated cadmium sulfate to which

0.015 M sulfuric acid was added had the same emftemperature coefficient as the neutral cell, while they

later [16] stated that the change in emf between 25

and 35 °C for some cells with electrolytes containing

0.01 mol sulfuric acid per liter of solution is slightly

less than that of neutral cells. Also later, Vosburgb

[17] after measuring the emf of acid cells from 15 to 40 °C stated that "Wolffs temperature formula was

found to hold for all [cells], the presence of acid being

found to have little or no effect on the temperature

coefficient." Vosburgh had used cells with sulfuric

acid concentrations of 0.021, 0.035 , 0.048, 0.064, and

0.078 mol per liter. Also the National Physical Laboratory [18] stated that the temperature coefficients of

standard cells of acidities from 0.05 to 0.5 N show no

particular correlation with the acidity.

At about the same time this latter work was done, the

Electrotechnical Laboratory of Japan [12] also investi- gated the effect of acid on the emf-temperature coeffi- cient of saturated standard cells; acidities ranged from

0.008 to 1.0 N and the temperature range was 15 to 30 °C. Ishibashi and Ishizaki [12] who conducted the

investigation found that the emf-temperature coeffi- cients of cells with electrolytes of low acidity were

irregular and some of them were smaller than those

for neutral cells. They also stated that "the cells acidified up to 0.08 mol/liter (0.16 AO have practically the same temperature coefficient of emf as given by Wolff

and Waters' equation, so far as the accuracy of only

0.005 percent is required [i.e., the same as neutral

cells]. The coefficient for separate cells, however,

must be determined whenever the accuracy of about

0.001 percent is required."

Vigoureux and Watts [19] also measured the emf

of "acid" cells as a function of temperature over the

range of — 20 to 40 °C. The solvent for the electrolyte, g CdS04

• — H2 0, in these cells was decinormal sulfuric

acid; therefore, the cell acidity was 0.077 A'. Vigoureux

and Watts did not, however, give comparisons with

"neutral" cells except to cite reference [18] to the

effect that the acidity of the electrolyte does not

affect the temperature coefficient to an appreciable

extent.

2. Experimental Detail

Owing to the various results reported above on the

emf-temperature coefficients of "acid" standard

cells of the saturated cadmium sulfate type and the

increased use of "acid" cells it was decided to make

new measurements of the emf-temperature coefficient of the "acid" type of cell. Accordingly, 12 cells made

at NBS in December 1955 were selected for the study.

These cells were made with 10 percent cadmium

amalgam, electrolytic mercurous sulfate, and crystals

g

of CdSCu • ^ H2O having a diameter of about 2 to 4

mm. The saturated solution of cadmium sulfate was

acidified to 0.021 N by the addition of sulfuric acid. The cells were of the H-form and were assembled as outlined in reference [8]; see also references [20] and

[21] for descriptions of the preparations of saturated

standard cells at the National Bureau of Standards.

These cells were also used in 1966 in the transfer of the "volt standard" from the old NBS site in Washington to the new NBS site in Gaithersburg, Md.

[22, 23]. In addition two different manufacturers of standard

cells each supplied 12 "acid" saturated standard cells for the study. These are subsequently referred to as commercial 1 cells or Cl cells and as commercial 2

cells or C2 cells. These cells were about 2 years old

at the start of the study. They were also of the H-form

and were made with 10 percent cadmium amalgam

and electrolytic mercurous sulfate. The Cl cells had a

stated acidity of 0.061 N (except cell C1B which was

replaced by one of acidity of 0.044 N for the measure- ments in Gaithersburg, Md.; the former one of acidity of 0.061 N was broken in transit from Washington,

D.C. to Gaithersburg, Md.) and were made with g

crystals of CdSG"4 x H2O of 4, 8, and 10 mm in diameter. The C2 cells had a stated acidity of 0.025 N and

g

the crystals of CdS04

• ~ H2O over the mercurous

sulfate paste had a consistency of granulated sugar

while those over the amalgam consisted of a mixture

of crystals of the consistency of granulated sugar and

of larger crystals about 8 mm in diameter. The depth

of mercurous paste was about 1.3 cm in NBS cells, 1.5 cm in Cl cells, and 1.7 cm in C2 cells. Studies of the effect of temperature on the emf

of these 36 cells were started in December 1961. The cells were housed in an oil bath maintained within

±0.001 °C at each of the temperatures used in the

study on the day measurements of the emf were made.

The temperatures were read with a calibrated Pt- resistance thermometer and the emfs were measured

to 0.1 /jlY with a Brooks comparator [24]. After a change

in temperature the cells were maintained at least one week at the new temperature before readings of the emf were taken. At least ten readings over a period

of two weeks were made at each temperature. The

sequence of the temperatures, the groups of cells used

at each temperature, and the time interval from the

termination of the last of the initial readings at 28

°C to the time of the last readings at subsequent

temperatures are given in table 1. Various interruptions, most notably the move of NBS

from Washington, D.C. to Gaithersburg, Md., in 1966,

delayed completion of the study until September,

265
TABLE 1. Sequence of measurements

Spmipnpp t\i Time intervals (days)* for NBS

cells CI cells C2 cells

W ashington, D.C.

28 °C NBS, CI, C2 0 0 0 32 °C NRS CI C2 42 59 44

35 °C NBS, CI, C2 108 105 108 38 °C NBS, CI, C2 147 147 147

rvirsb, ci. 190 197 190 35 °C NBS, CI. C2 253 266 265 28 °C NBS, CI, C2 351 352 351 20 °C NBS, CI, C2 506 507 506 20 °C NBS, CI. C2 625 629 628

25 °C NBS. CI. C2 741 742 741

Gaithersburg, Md.

15 °C CI. C2 1938 1937 25 °C NBS 2246

20 °C

15 °C NBSNBS

2303 2414

*From final date at one temperature to final date at subsequent

temperature.

1968. Owing to the time period, about 7 years, con- sumed in the study, the cells showed slight drifts in emf

with time. These were made evident by comparing

the mean emf of the 12 cells of each group at those

temperatures where second sets of measurements

were made. The average changes or drift rates were

found to be 3.62 X 10"9 V/day for the NBS cells,2 2.71 X10-9 V/day for the Cl cells, and 9.6 X lO10 V/day for the C2 cells.

First the observed emfs without drift-rate corrections were fitted to the third-order equation: 3

E, = E2o°c + <*{t - 20) + (3(t - 20) 2 + y(t - 20) 3

(2) by the method of least squares using a computer pro- gram: £20 °c is the emf at 20 °C and a, /3, and y are constants. The values of a, (3, and y for the NBS, Cl,

and C2 cells are given, respectively, in tables 2, 3,

and 4. The mean standard deviations are given for a, /3, and y and the uncertainties in the fit, lcr, are standard deviations of the fit of the data to equation

(2). Corrections for drifts in emf with time were then

made for each cell. These were obtained from the

equation:

£'-£ = A£ = a'(<-20)+/3'(*-20) 2 + y'(f-20) 3

(3) where E' is the corrected emf and a' — a+ A, f3' =

/3 + A, and y' = y+A with A being the correction to the constants of eq (2). The final values for the constants for each cell and the averages are given within

the parentheses in tables 2, 3, and 4. Final recommended values for a, ft, and y for each

type of cell are given at the bottom of tables 2, 3, and

4; data for cell C2J were omitted in arriving at a final value owing to the large standard deviation of the fit to eq (2). Thus, for NBS cells of 0.021 N acidity,

E, = E20 -c ~ [40.44(; - 20) + 0.921 (t - 20) 2 - 0.00866U - 20) 3

] x 10~6 V (4)

Table 2. Coefficients in the equation: E = E20 °c+ a(t — 20) + /3(t — 20) 2 + y(t — 20) 3 for National Bureau of Standards cells

Cell No. E20 °c V

(On Vas and tee scales)

amV

y cr mV

NBS1157 #a 1.0186569(4) *- 40.4096 (.01) *- 0.9271 (.008) *0.0077(. 00070) 1.3194 NBS1158 1.0186598(3) -40.4076(.01) -0.9498 (.003) .0086(. 00061) 1.2891 NBS1159 1.0186597(3) -40.4971 (.01) -0.9304 (.007) .0080 (.00069) 1.3941 NBS1160 1.0186599(4) -40.4061 (.01) - 0.9260(.005) .0077 (.00063) 1.0890 NBS1161 1.0186619(6) -40.7635(.02) -0.9221 (.010) .0080(. 00136) 1.9925 NBS1162 1.0186507(3) - 40.2162 (.01) -0.9216(.003) .0075(. 00060) 1.0133 NBS1163 1.0186528(2) -40.3923 (.01) -0.9194(.002) .0077(. 00054) 1.4056 NBS1164 1.0186580(6) -40.6369(.01) -0.9221 (.008) .0081 (.00099) 1.7193 NBS1165 1.0186529(4) -40.6680(.01) -0.8893 (.008) .0069(. 00070) 1.5041 NBS1166 1.0186549(1) - 40.7567(0) - 0.8840(0) .0067(.00032) 1.3328 NBS1167 1.0186602(6) - 40.1705(.01) - 1.0128(.009) .0114(.00104) 2.3725 NBS1168 1.0186537(2) -40.1258(0) - 0.9170(.001) .0071 (.00035) 0.5679 mean 1.0186564(4) - 40.4542 (.01) -0.9268(.005) .0080(. 00071)

0.0630 0.0093 .00035 1.4847

final -40.44 - 0.921 0.00866 1.5 # Values within parentheses are negative.

* All values within parentheses are positive corrections for drift in emf with time.

3 Values in parentheses for this column are corrections for the last decimals in the emf or should be multiplied by 10-7

.

2 Earlier observations, from December 1955 to December 1961 also gave this drift rate 3 Inspections showed that a second-order equation was inadequate whereas a fourth'

for these cells. order equation gave no better fit to the experimental data than a third-order equation. 266
Table 3. Coefficients in the equation: E = E20 «c + a (t — 20) + /3 (t — 20f +y (t— 20)3 /or Company 1 cells

Cell No. E20 • v

(On V69 and f68 scales)

a

(J, „ VV pM

,, vv y

„ v 11V CIA # a 1.0186005(8) *— dn4'ly. 11 ldjurn ItO \ \> ) *_ n Ron? 1 rifli 1 *0U.UW1 ftfVil r OfMVlRI ^ . vWVO^ n d.od.7 C1B 1.0185969(24) -40.5868(0) -0.8039 (.001) .0031 (.00033) .4359 C1C 1.0185995(27) -40. 1137 (.01) -0.8963 (.001) .0064(. 00035) .6385 C1D 1.0186021(43) -40.2768(.01) -0.8933 (.003) .0065 (.00050) .8085 C1E 1.0185992(45) -40.4774(.01) -0.8608(.003) .0055(.00054) .6251 C1F 1.0186026(51) -40.3271 (.01) -0.9174(.004) .0073 (.00058) .7664 C1G 1.0186006(32) -40.0042 (.01) -0.9063 (.001) .0067 (.00042) .2867 C1H 1.0186011(61) -40.1528(.01) -0.8850(.007) .0060(. 00059) .3872 C1I 1.0186007(40) -39.9425 (.01) -0.9064 (.002) .0067(. 00050) .4037 C1J 1.0186000(32) -39.8191 (.01) -0.9244(.001) .0072 (.00042) .7741 C1K 1.0185987(37) -39.9253 (.01) -0.9095 (.002) .0067 (.00046) .7795 C1L 1.0185999(67) -40.0702 (.01) -0.9015(.008) .0067 (.00062) .5839 mean 1.0186002(38) -40. 1509 (.01) -0.8913(.003) .0062 (.00044)

crm 0.0662 0.0093 0.00029 0.6067

final -40.14 -0.888 0.00668 0.6

* Values within parentheses are negative

*AU values within parentheses are positive corrections for drift in emf with time.

a Values in parentheses for this column are corrections for the last decimals in the emf or should be multiplied by 10-7

.

Table 4. Coefficients in the equation: E= E20 °c+a (t — 20) +/3 (t — 20f + y (t — 20ffor Company 2 cells

Cell No. £20 »c V

(On and t6s scales)

a MV

y

p.\

cr (l\

C2A a 1.0186219(52) *- 38.8723 (.01) -1.0420(.005) 0.0109 (.00060) 1.9416 C2B #1.0186231(21) -39.6109(0) -0.9413 (.001) .0075 (.00031) 1.1653 C2C 1.0186221(4) -39.4001(0) -0.9683(0) .0084(. 00007) 1.5217 C2D #1.0186220(10) -40.2280(0) -0.8616(0) #.0051 (.00018) 0.8837 C2E 1.0186223(31) -38.9803 (.01) -1.0480 (.001) .0112(.00045) 1.7775 C2F 1.0186214(31) -39.3767(.01) -0.9904 (.001) .0095(. 00045) 1.3334 C2G 1.0186254(15) -39.2822(0) -0.9942 (.001) .0094(. 00020) 1.6330 C2H 1.0186199(23) -38.9568(.01) -1.0350(.001) .0106(. 00031) 1.8220 C2I #1.0186169(42) -40.2187(.01) -0.8390(.003) .0040(.00053) 2.0724 C2J #1.0186041(368) #-39.9149(1.45) #-0.7728(.240) .0008 (.00950) 7.2131 C2K 1.0186217(46) -38.5832(.01) -1.0853 (.004) .0124(. 00054) 2.5660 C2L #1.0186218(4) -38.7256(0) -1.0649(0) #.0115(.OOO07) 2.3618 mean #1.0186202(20) #-39.3458 (.12) #-0.9702 (.019) .0084(.00035)

<Tm 0.1669 0.0262 .00101 2.7024 mean-C2J 1.0186217(11) -39.2941 (.01) -0.9882 (.002) .0091 (.00029)

(Tm 0.1669 0.0243 .00081 1.7991

final -39.28 -0.986 .00943 1.8

# Values within parentheses are negative.

*A11 values within parentheses are positive corrections for drift in emf with time unless noted.

a Values in parentheses for this column are corrections for the last decimals in the emf or should be multiplied by 10-'.

for CI cells of 0.061 N acidity,

£( = £2 o°c - [40.14(r-20) + 0.888(t-20) 2 - 0.00668(t - 20) 3 ] x 10-«V (5)

for C2 cells of 0.025 N acidity,

Et = E29 .c - [39.28(t-20) + 0.986(f - 20) 2 - 0.00943 (t - 20) 3

] x 10-«V. (6) At about the same time covered by this investigation,

Froehlich and Melchert, with the assistance of Steiner,

in the laboratories of the Physikalisch-Technische

Bundesanstalt, also carried out a study of the emftemperature coefficient of "acid" standard cells over

the range of 10 to 30 °C and published their results

in April, 1971 [25]. The pH of the electrolyte used

in the cells was 1.4±0.2; the hydrogen-ion activity was, therefore, of the order of 0.04. No concentration

value was given; from known values of the activity coefficient of saturated cadmium sulfate solution,

namely, 0.03641 [1] and of sulfuric acid of the same

ionic strength, namely, 0.208 [26], and the density of the saturated solution [27] the normality calculated

from the pH value given by Froehlich and Melchert

would appear to be about 0.26 N. However, the emfs

given for 20 °C by Froehlich and Melchert indicate

267
that the acidity was about 0.03-0.04 N. Ten cells were

used in the study, two each of about three, four, five, seven, and nine years old at the time of the measurements. The emfs at 20 °C ranged from 1.0186053 V

to 1.0186457 V; no correlation with pH was given,

however, a relation between E and dEjdT may be

calculated from their data and is shown later. From the results obtained for the ten cells over

a temperature range of 10 to 30 °C and taking cog- nizance of drifts in emf with time, Froehlich and

Melchert [25] arrived at: Et = E20 °c - [39.78(^-20) + 0.936(^-20) 2 -0.0086U-20) 3

] x 10-6 V (7)

for the relation between emf and temperature.

In table 5, values for dE/dT from 15 to 40 °C are given for NBS, Cl, and C2 cells. The standard devia-.

tion, a, of an individual value and the standard

deviation of the mean, cr m , are also given. The values

of dE/dT given by ihe international formula (Int), the equation of Vigoureux and Watts (VW), and of Froehlich and Melchert (FM) are given for comparison.

Also the mean emfs are given at the bottom of the table. Inspection of the data shows that there is no correlation between dE/dT and the normality of the solution. This is clearly shown in figure 1 for values of dE/dT

at 20 °C; similar results are found for the other temperatures. It is evident, as was stated earlier by

Ishibashi and Ishizaki [12] that the emf-temperature

coefficients of cells with electrolytes of low acidity are irregular. Since the emf is decreased by the

addition of acid it remains that dE/dT may be a function

of the emf. This is shown not to be the case in figure 2 where dE/dT at 20 °C is plotted against the emf at 20 °C; 10.5 /i,V (see ref. [8] for this factor) were subtracted from the emfs of Ishibashi and Ishizaki to place their emfs on the United States of America

base of reference for the volt. It is evident from figure 2 that there is no clear correlation between E and

dE/dT over the acidity range of 0 to 0.06 yV. As also stated earlier by Ishibashi and Ishizaki for a precision

of 0.005 percent the addition of acid has little or no

effect on dE/dT while for a precision of 0.001 percent

the individual values of dE/dT must be experimentally

determined. Furthermore, drift rates must be deter- mined if the highest precision is required.

One point is significant in figure 2, namely, that some of the emf data of NBS and ETL at 20 °C exceed

that normally attributed to "neutral" cells and that some of the emfs reported by Froehlich and Melchert

are very close to that for the so-called "neutral" cell. The international value of the emf of "neutral" cells was based on the measurements made with cells with chemically prepared and with dc-electrolytic mercurous sulfate. Wolff and Waters [2] showed that dc-electrolytic mercurous sulfate, on the average, gave cells with emfs exceeding by 16.8 (x\ those made

with chemically prepared mercurous sulfate. Since

dc-electrolytic mercurous sulfate was used in the cells discussed herein, this value of 16.8 /aV was added to the international value and is designated as "corrected

emf in figure 2. With this correction the emfs of the

NBS cells approximate that of the so-called "neutral"

cell; apparently, the acid in the cells is largely neutral- ized by the alkaline constituent of the glass container

during the aging period (this is exhibited by the upward

drift in emf with time, as mentioned earlier in this paper).

Finally in table 6 values of the second derivative,

d

2E/dT2

, and in table 7 values of the entropy and heat- capacity changes for the cell reaction in "acid"

for 20 °C

(DEI <D

•Internationa!

a = activity H*

FIGURE 1. Emf-temperature coefficients of saturated standard cells and the normality

ofsulfuric acid in the electrolyte in the cell. O—National Bureau of Standards [1]. ©—National Physical Laboratory [18]. 0—Eleclrotechnical Laboratory, Tokyo [12]. x —PhysikaJischTechnische BundesanstaJt [25], O—Vigoureux and Watts [19].

B—Obata and Ishibashi [14]. ©—Company 1. O—Company 2. ©—International formula [2,3,4].

268
TABLE 5. Emf-temperature coefficients of "acid" standard

OA O/"1 zu cor °C ZD

\I« *

INO. U 1 i^9 IN 155 C 9 C 1 c 1 C2

i1

0.A AA — oU. * o — z / .Oo Af\ A<r\ /in i i — QQ OA 00.00 A Q CA A Q Q7

04QA 0/1 QQ OA on AO A(\ Af\ Af\ CQ QQ Al AQ 1 7 AO Of. A Q A 0. Qo OA AH OA (LA OQ AO AA AQ AA 1 A on /in AQ A7 — AQ Qc: 4-0.00 — AQ ACL '*O. i+0

A 1C\ £A OA Q/l AA AC\ Aa 97 AA 9Q AO AO AO A/1 AO AO

5 - 30.92 -31.44 -27.63 - 40.74 - 40.47 - 38.97 -49.16 - 48.59 - 48.57 6 — 30.41 — 30.59 — 28.73 — 40.21 — 40.32 — 39.37 — 48. 78 — 48.86 — 48.51

7 - 30.59 - 30.41 - 28.63 - 40.38 - 39.99 -39.28 - 48.94 -48.51 - 48.49 8 - 30.80 - 30.87 — 27.79 — 40.63 - 40.14 — 38.95 — 49.09 -48.43 - 48.47

9 - 31.28 - 30.35 -31.51 - 40.66 - 39.93 -40.21 — 48.90 - 48.44 - 48.23 10 - 31.39 - 30.00 c (- 30.46) - 40.76 - 39.81 f (- 41.36) - 49.07 - 48.47 c (- 50.72)

11 -29.19 - 30.30 - 26.79 - 40.16 - 39.92 - 38.57 - 49.27 - 48.45 - 48.42

12 - 30.41 - 30.58 -27.22 -40.13 - 40.06 - 38.73 -48.73 - 48.45 - 48.52 mean - 30.58 - 30.76 c- 28.72 - 40.45 -40.14 <•- 39.29 - 49.01 - 48.52 <•- 48.45

<T 0.557 0.410 1.559 0.216 0.232 0.554 0.159 0.092 0.090

0"m .161 .118 .470 .062 .067 .167 .046 .027 .027

Int. "-m.Z9(.{A) -40.64(.04) -40.39(.04)

VW "-29. 79 (.71) -39.40(.55) -48.02(.55)

FM "-29.78(.06) -39.78(.05) -48.50 (.03)

E(NBS) 1.0188353 1.0186572 1.0184331 E(C1) 1.0187817 1.0186040 1.0183819 E(C2) c 1.0187934 c 1.0186228 c 1.0184029 FM 1.0187982 1.0186238 1.0184026

Int. 1.0188224 1.0186442 1.0184185

* Numbers are in sequence for cells NBS, CI, and C2 given in tables 2, 3, and 4.

" Values in parentheses standard deviation of the mean.

6 Maximum uncertainty of deviations from given formula.

r Eleven cells only; data for cell No. 10 omitted.

d Vigoureux and Watts gave only the differences between emfs at a particular temperature and emfs of reference cells at 20 °C.

Table 6. The second derivative, d

2 E/dT2

15 °C 20 °C 25 °C

No.* NBS CI C2 NBS CI C2 NBS CI C2

~ 1," -2.090 - 1.964 - 2.419 - 1.838 - 1.778 - 2.074 - 1.586 - 1.593 - 1.729

2 -2.170 - 1.711 - 2.115 - 1.894 - 1.606 - 1.881 - 1.617 - 1.501 - 1.646

3 -2.108 - 1.993 - 2.191 - 1.847 - 1.791 -1.937 - 1.586 - 1.588 - 1.683 4 -2.092 - 1.991 - 1.871 - 1.842 - 1.781 - 1.723 - 1.592 - 1.571 - 1.676

5 -2.105 - 1.897 - 2.444 - 1.824 - 1.716 - 2.094 - 1.543 - 1.534 - 1.745

6 - 2.080 - 2.063 - 2.277 - 1.837 - 1.827 - 1.979 - 1.594 - 1.590 - 1.680

7 - 2.082 - 2.024 - 2.274 - 1.835 - 1.811 - 1.986 - 1.588 - 1.597 - 1.698

8 - 2.101 - 1.954 - 2.395 - 1.828 - 1.756 - 2.068 - 1.556 - 1.558 - 1.741 9 - 1.991 - 2.025 - 1.808 - 1.763 - 1.809 - 1.672 - 1.535 - 1.593 - 1.536

10 - 1.979 - 2.075 (-2.335) - 1.768 - 1.847 (- 2.026) - 1.557 - 1.618 (- 1.717)

11 -2.381 - 2.030 - 2.551 - 2.008 - 1.815 -2.163 - 1.634 - 1.600 - 1.774

12 - 2.056 -2.004 - 2.473 - 1.832 - 1.787 - 2.130 - 1.509 - 1.570 - 1.787 mean -2.103 - 1.978 c - 2.256 - 1.843 - 1.777 c - 1.973 - 1.575 - 1.576 c - 1.709

(T 0.088 0.096 0.212 0.058 0.064 0.159 0.036 0.031 0.070

<Tm .025 .028 .067 .017 .019 .048 .010 .009 .021

Int. a -2.200(.005) - 1.900(.003) - 1.600(.003) VW b -2.049(.08) - 1.806(.06) - 1.653(.06) FM -2.130(.007) - 1.872(.004) - 1.614(.007)

*Numbers are in sequence for cells NBS, CI, and C2 given in tables 2, 3, and 4.

a Values in parentheses are am , standard deviation of the mean.

b Maximum uncertainty of deviations from given formula.

c Eleven cells only; data for cell No. 10 omitted. 269
cells of the saturated cadmium sulfate type, jU.V/°C

30 °C 35 °C 40 °C NBS CI C2 NBS CI C2 NBS CI C2 - 56.28 -56.04 -56.15 - 62.32 - 62.62 -62.21 -67.10 -68.27 - 66.54 - 56.57 - 55.62 - 56.07 -62.58 - 62.36 - 62.55 - 67.22 - 68.59 - 67.85 - 56.35 - 55.98 - 56.23 -62.32 - 62.41 -62.73 - 67.00 - 67.82 - 67.97 - 56.32 - 55.97 - 55.98 -62.40 - 62.25 - 62.76 - 67.24 -67.48 - 68.79 -56.18 - 55.81 - 56.42 -61.79 »• 62.12 - 62.52 -66.00 - 67.53 - 66.87 -56.15 - 56.22 -56.17 - 62.30 - 62.40 - 62.33 - 67.23 - 67.40 -67.00 - 56.08 - 55.96 - 56.27 - 62.34 -62.35 - 62.60 -67.19 - 67.66 - 67.49 -56.18 -55.73 - 56.35 -61.91 - 62.03 - 62.60 -66.28 -67.35 -67.21 - 56.00 - 55.86 -55.57 -61.97 - 62.20 - 62.23 -66.79 - 67.47 - 68.21

C.A 11 — oo.yy \ — oo.oo ) AO AO 37 Ci C.A Qf\\ — A7 AO o ( .oy — A7 AO Oi.OU cf AO c:o\ (— W.bZ) - 56.50 - 55.92 -56.32 -61.88 -62.31 -62.28 -66.75 - 67.62 - 66.30 -56.21 - 55.76 - 56.59 -62.58 -61.98 - 62.96 -67.83 -67.12 -67.61 - 56.26 - 55.91 c- 56.19 - 62.24 - 62.28 c- 62.52 - 67.03 - 67.66 c- 67.44

0.165 0.160 0.262 0.284 0.179 0.241 0.521 0.405 0.688

.048 .046 -56.63(.04)

<'-56.10(.63) - 55.92

.079 .082 .052 -62.37(.06)

<^64.08(.70)

.073 .150 .117 -66.60(.09)

e- 72.41 (.70)

.297

Electromotive

forces d 1.0181694

1.0181205

c 1.0181408

1.0181410

1.0181528

1.0178726

1.0178246

c 1.0178436

1.0178547

1.0175493

1.0174994

c 1.0175182

1.0175317

e Based on four cells only; Vigoureux and Watts considered their results above 28 °C to be less reliable than those for lower temperatures.

Int. International formula [2, 3, 4]. VW Vigoureux and Watts [19]. FM Froehlich and Melchert [25].

for saturated standard cells, /xV/°C/°C

30 °C 35 °C 40 °C NBS CI C2 NBS CI C2 NBS CI C2 - 1.334 - 1.408 - 1.384 - 1.082 - 1.222 - 1.039 -0.830 - 1.037 - 0.694 - 1.341 - 1.3% - 1.412 - 1.065 - 1.291 - 1.078 -0.788 - 1.186 -0.943 - 1.325 - 1.386 - 1.428 - 1.065 - 1.183 - 1.174 -0.804 - 0.981 - 0.920 - 1.342 - 1.361 - 1.428 - 1.092 - 1.151 - 1.280 -0.842 - 0.941 - 1.133 - 1.263 - 1.353 - 1.395 - 0.982 - 1.172 - 1.046 - 0.701 -0.991 - 0.696 - 1.351 - 1.354 - 1.381 - 1.108 - 1.118 - 1.083 - 0.865 -0.881 -0.784 - 1.340 - 1.383 - 1.410 - 1.093 - 1.170 - 1.122 -0.846 - 0.956 - 0.834 - 1.283 - 1.361 - 1.413 - 1.010 - 1.163 - 1.086 - 0.737 -0.965 - 0.759 - 1.307 - 1.377 - 1.400 - 1.079 - 1.161 - 1.264 -0.851 -0.945 - 1.128 - 1.347 - 1.390 (- 1.408) - 1.136 - 1.161 (- 1.099) - 0.926 -0.932 (- 0.790) - 1.261 - 1.385 - 1.386 -0.888 - 1.171 -0.998 -0.515 - 0.956 - 0.610 - 1.385 - 1.353 - 1.444 - 1.162 - 1.136 - 1.101 - 0.938 -0.919 - 0.758 - 1.323 - 1.392 - 1.408 - 1.072 - 1.177 - 1.116 -0.804 - 0.974 -0.842

0.038 0.080 0.020 0.068 0.044 0.105 0.113 0.077 0.172

.011 .023 - 1.300(.003) - 1.590(.07).006 .020 .013 - 1.000(.005)

d - 1.617(.08)

.032 .033 .022 -0.700(.008)

d - 1.734(.09)

.052

- 1.356(.013)

d Vigoureux and Watts gave only differences between emfs at a particular temperature and emf of reference cells at 20 °C.

Int. International formula [2, 3, 4]. VW Vigoureux and Watts [19]. FM Froehlich and Melchert [25]. 270
TABLE 7. Entropy and heat-capacity changes for the reaction in "acid" standard cells ofthe saturated cadimum sulfate type

Entropy changes, J K~' moh1

Acidity N 15 °C 20 °C 25 °C 30 °C 35 °C 40 °C

International [2,3,4] N3S

FM

C2

CI vw

Thermochemical [1]

International [2.3,4] NBS

FM

C2

CI VW

Thermochemical [1]

0.00092

.021

* .025

.061 .077 0

-5.86 -5.90

-5.75 -5.54

-5.94

-5.75

-7.84

-7.81 -7.68

-7.58

-7.75 -7.60

-9.53 -9.46

-9.36

-9.35 -9.36

-9.27 -9.58

-10.93 -10.86

-10.79

-10.84 -10.79

"(-10.83)

-12.04

-12.01 -12.85 -12.94

-12.06

-12.02

"(-12.37)

-13.01 -13.06

"(-13.97)

Heat-capacity changes, J K

_1 mol-1

0.00092

.021

* .025

.061 .077 0

-122.33 -116.94

-118.44

-125.45 -109.99

-113.94

-107.48

-104.26 - 105.90 -111.71 -101.64

—102.17

-92.06

-90.62 -92.86

-98.33 -90.68

-95.10

-93.72

-76.05 -77.40

-79.33 -82.37

-81.43

"(-93.02)

-59.46

-63.75

-42.30 -48.59

-66.36

-69.99

"(-96.15)'

-50.88 -58.%

"(-104.79)

* pH=1.4±0.2; aH+ = 0.04.

" Based on four cells only [19].

International — International formula [2,3,4]. FM Froehlich and Melchert [25]. VW Vigoureux and Watts [19].

I.OI8) /jV

FIGURE 2. Emf-temperature coefficients of saturated standard cells and the emfof the cell. O—National Bureau of Standards [11 ©—Electrotechnical Laboratory, Tokyo [12]. X —PhysikaBsch-Technische Bundesanstalt [25]. ©—Company 1. ©—Company 2. •—International formula [2,3,4]. —Corrected international emf.

standard cells are given. The entropy change for the

"acid" cells is in all cases less than those observed

for the so-called "neutral" cell while the heat-capacity

change for "acid" cells relative to the "neutral" cell

is less well defined.

3. References

[1] Hamer, W. J., J. Res. Nat. Bur. Stand. (U.S.), 76A (Phys. and

Chem.), No. 3, 185-205 (May-June 1972).

[2] Wolff,W. A., and Waters, C. E. , Bull BS 4, 70 (1907).

[3] Wolff, W. A., Bull. BS 5, 309 (1908).

[4] International conference on electrical units and standards, Appendix to report: Notes to the specifications as to the methods adopted in various standardizing laboratories to realize the international ohm and the international ampere, and to prepare the Weston Normal Cell (Darling & Sons,

Ltd., London, 1908).

[5] Guoy, M., Compt. rendu. 130, 1399 (1900).

[6] Hager, O. B., and Hulett, G. A., J. Phys. Chem. 36, 2095 (1932).

[7] Craig, D. N., and Vinal, G. W., J. Res. NBS 1 7, 709 (1936)RP939.

[8] Hamer, W. J., Standard cells, their construction, maintenance, and characteristics, NBS Monograph 84, 1965.

[9] Smith, F. E. (Sir Frank), A Dictionary of Applied Physics, p. 268, Ed. Sir Richard Glazabrook, (Macmillan and Co.. Ltd., London, 1911).

[10] Report of Natl. Phys. Lab., Electrician 75, 463 (1915).

[11] Obata, J., Proc. Math. Phys. Soc. (Japan), (3), 2, 232 (1920).

[12] Ishibashi, V., and Ishizaki, T., Researches Electrotechnical

Lab.. No. 318 (1931).

[13] Vosburgh, W. C.,J. Am. Chem. Soc. 47, 1255(1925).

[14] Obata, J., and Ishibashi, V., Researches Electrotechnical Lab., No. 88 (1921).

[15] Vosburgh, W. C, and Eppley, M., J. Am. Chem. Soc. 45 , 2268

(1923). [16] Vosburgh, W. C, and Eppley, M., J. Am. Chem. Soc. 46, 110

(1924). [17] Vosburgh, W. C, J. Opt. Soc. & Rev. Sci. Instrument 12, 511

(1926).

[18] Rapport, Comite Consultatif d'Electricite Proces-Verbaux des Seances de 1930. in Proces-Verbaux des Seances, Comite In- ternational des Poids et Mesures Deuxieme Serie-Tome XIV,

p. 171,1931.

[19] Vigoureux, P., and Watts, S., Proc. Phys. Soc. (London) 45,

172(1933).

[20] Brickwedde, L. H., Compt. rendus Quatorzieme Conference IUPAC, p. 105. London, 1947.

[21] Hamer, W. J., and Law, C. A., Comite Consultatif d'Electricite, ll e Ses, Doc. No. 5, May 10-12, 1965.

[22] Hamer, W. J., J. Wash. Acad. Sci. 56, 101 (1966).

[23] Hamer, W. J., Inst. Soc. Am., M2-l-Mestind-67, 1967.

[24] Brooks. H.B..BS J. Res. 11,211 (1933).

[25] Froehlich, M., and Melchert, F., with Steiner, O., Metrologia 7, No. 2,58(1971).

[26] Harned, H. S., and Hamer. W. J., J. Am. Chem. Soc. 57, 27

(1935).

[27] International critical tables 3, 66 (1928). (Paper 76A4-723)

271
JOURNAL OF RESEARCH of the National Bureau of Standards — A. Physics and Chemistry

Vol. 76A, No. 3, May-June 1972 Thermodynamics of Standard Cells of the Saturated

Cadmium Sulfate Type

Walter J. Hamer

Institute for Basic Standards, National Bureau of Standards, Washington, D.C. 20234

(December 21, 1971)

This paper gives data on the thermodynamic functions of standard cells of the saturated cad- mium sulfate type, as obtained from calorimetric and equilibrium data at 25 °C or from the electro- motive forces (emfs) and emf-temperature coefficients of the cell for the temperature range of 0 to 43.6 °C. The functions considered are the changes in Gibbs energy, enthalpy, entropy, and heat capacity

for the cell reaction. The electromotive forces are expressed on the Ve» volt and the ten temperature

scale. Results are expressed on the SI and for comparisons with literature data in terms of the defined thermochemical calorie. The effect of expressing the emf-temperature coefficient as a function of temperature in different ways on the values for the changes in entropy and heat capacity for the cell reaction is discussed. Finally, the observed emf of the standard cell at 25 °C is compared with emfs

calculated from various values reported for the standard potentials of the cadmium-amalgam and mer- cury-mercurous sulfate electrodes and the activity coefficient of cadmium sulfate in saturated aqueous

solution.

Key words: Emf-temperature coefficients of standard cells; emfs of standard cells; standard potential

of standard cells; thermodynamics of standard cells.

1. Introduction

It is the purpose of this paper to present data on the

thermodynamic functions of standard cells of the sat- urated cadmium sulfate type,1 as obtained from calori- metric and chemical equilibrium data, and to compare

these data with those obtained from the electromotive

force (emf) and emf-temperature coefficient of the cell. The thermodynamic functions considered are the

changes in Gibbs energy (free energy), AG; enthalpy

(heat content), AH; entropy, AS; and heat capacity at constant pressure, ACP, where the changes are equal

to the differences between the thermodynamic quantities of the products and the reactants of the cell re- action. Thermodynamically, these are given or are

related by:

AG = AH-TAS = AH + T[d(AG)ldT] p (1)

AS ACDd In T

and

ACP = [d(AH)ldT] p

(2)

(3)

where T is the Kelvin temperature (defined in the ther- modynamic scale by assigning 273.16 K to the triple

point of water; freezing point of water, 273.15 K [l] 2

). The thermodynamic functions for a galvanic cell (or

standard cell) may be obtained from its emf, E, and its emf-temperature coefficient, dEjdT, and are given by:

AG - - nEF (4)

A// — — nEF + nFT(dEldT) (5)

and

AS — nF( dE/dT

)

ACP = nFT(d'2E/dT2

)

(6)

(7)

where n is the number of equivalents involved in the

cell reaction (in the present case n is 2) and F is the

Faraday. F has a value of 96487 coulombs per gramequivalent [1, 2, 3, 4] or 23060.9 calories per volt per

gram-equivalent.3 Accordingly, a comparison may be

made between the thermodynamics ot the overall re- action of a standard cell as determined from calori- metric and chemical equilibrium data or from emf data.

1 Also referred to as the Weston cell after Edward Weston who invented the cell in 1892.

2 Figures in brackets indicate the literature references at the end of this paper.

3 Based on the relation 1 thermochemical calorie (defined)^ 4.1840 J [1].

272
2. The Standard Cell (saturated CdS04 type)

The conventional standard cell of the saturated

cadmium sulfate type,4 considered here, may be

represented by:

(-)Cd,Hg(2p) CdS04 -^H2 0(s)

8 H2 0(s)

CdS04 (ss)

Hg2 S04 (s)

CdS04

Hg(l)(+)

where the vertical lines indicate the interface between

two distinct phases (the amalgam consists of 2 phases, one solid and one liquid, but for simplicity vertical

lines are not used in designating it), 2p = 2 phases,

ss = saturated solution, s = solid (or paste), and

Illiquid. The cell reaction is [5]:

*Cd,yHg(2p) + Hg2 S04 (s) +

•mH2 0(ss)

8/3

to -8/3

to

CdS04

to -8/3

CdS04 (8)

•-H2 0(s)+2Hg(l) + U-l)Cd,yHg(2p)

where x moles of Cd are associated with y moles of

Hg in the amalgam and to is the number of moles of water associated with 1 mole of CdS04 in the satu- rated solution. Ten percent amalgams are now generally used, although there is now a trend to a return

to the use of 12|-percent amalgams as they are more

suitable for portable (or shippable) cells. At 25 °C to has a value of 15.089 [6]. The overall reaction for the cell may be considered

as the sum of five reactions, namely:

(a) Cd(s)+Hg2 S04 (s)^CdS04 (s)+2Hg(l) (9)

(b) CdS04 (s)+H2 0(l)^CdS04 -H2 0(s) (10)

(c) CdS04 -H2 0(s)+|H2 0(l)^CdS04 -|h2 0(s)

(d)

and

(e)

CdS04

• toH2 0(ss) ->! H2 0(1)

to — 8/3 3

+ -^rCdS04 -f H2 0(s)

to — 8/3 3

Cd(10% amalgam)^ Cd(s)

(11)

(12)

(13)

for which, reaction (e) excepted, calorimetric or thermal data are available.

3. Calorimetric and Chemical

Equilibrium Data

Reaction (a)

For reaction (a) Cohen, Helderman, and Moesveld

[7] obtained -45.346 cal mob 1 at 18 °C for the differ- ence between the heats of formation of CdS04 and

Hg2 S04 ; this difference is also equivalent to the heat

of reaction (a) since the heats of formation of the ele- ments, Cd and Hg, in their standard states are zero. 5 However. Cohen et al. arrived at their value for the heat

of reaction (a) from combined calorimetric and emf

data. They obtained their value by combining the value

for the heat of solution of CdS04 in water at 18 °C to give a solution 2.559 percent in CdS0 4 with the value

for the heat of reaction of an unsaturated standard cell (2.559% in CdS04 ) which they obtained from the ob- served emf of the cell at 18 °C and the measured emftemperature coefficient of the cell at 18 °C. For our purpose, however, we need to obtain the heat

of reaction (a) directly from calorimetric data. This heat

value may be obtained from the heats of reaction of Cd

and CdS04 with other substances and of Hg and Hg2 S04 with other substances, as outlined by the following

equations (unfortunately, most of the data were ob- tained many years ago). In these equations, the numbers in parentheses indicate, according to convention,

the number of moles of water in v/hich the substance is dissolved. Although, according to convention, the moles

of water involved in dilution or solution [eqs (a4) and

(a5) under Cd and CdS04 , for example] are not included

in representing the reaction they are included here so

that the equations balance, but are shown in parentheses with an asterisk to indicate that the heat of formation of water is not used in determining the heat of dilu- tion or solution. Likewise, when the water contains KI

or KI + I 2 , it is also shown in parentheses with a superscript to indicate that the heat of formation of this water,

containing KI or KI +

1

2 , is not used in determining the

heat of dilution or solution (in the summation of the fol- lowing reactions this quantity cancels). Also, in each

case the values reported in calories were first con- verted to international joules using the factor used by

the individual authors; these were then converted to absolute joules using the factor 1.000167 and these in turn were then converted (for comparison purposes)

to calories using the presently accepted thermochemical definition of the calorie (see footnote 3) and are designated by the symbol cf meaning conversion factor. Also in the equations that follow, s = solid, 1 = liquid,

and g = gas. The numbers in parentheses after each

value for A// denote the ± uncertainty, discussed

later.

5 Calorie is retained for the unit of energy when the literature is cited. More is given on-

' This is the type of standard cell presently used in the maintenance of the unit of emf this point including the use of SI later in this paper. The thermochemical calorie is presently

in National Laboratories. / defined in terms of the joule by the relation given in footnote 3.

273
Cd and CdS04

(al) Cd(s) + 2 HC1(200) -» CdCl2 (400) + H2 (g)

A//, (20 °C) = - 17,230(70) cal mob1 = - 72,033(293) J mol"1 [8] A//, (18 °C) -- 17,290(90) cal mol" 1 = - 72,284(377) J mol"1 [8]

A//,(18°C) cf =- 17,276(90) cal mol"1 =- 72,284(377) J mol"1

(a2) CdCl2 (400) + BaS04(s) -» CdSO4(200) + BaCl2(200)

A//2 (19 °C) =5,683(55) cal mc-H = 23,742(230) J mol-' [9]

A//2 (18 °C) =5,793(60) cal mol 1 = 24,201(251) J mol 1 [10, 11]

A#2 (18 °C) cf =5,784(60) cal mol" 1 = 24,201(251) J mc-H

(a3) BaCl2(400) + H2SO4(400) -> BaS0 4(s) + 2 HC1(400)

Atf3 (19 °C) = - 9,152(55) cal mol"1 = - 38,234(230) J mol"1 [9]

A//3 (18 °C) = - 9,262(60) cal moH = - 38,694(251) J mol"1 [10, 11]

A//3 (18 °C) cf = - 9,248(60) cal mob1 = - 38,694(251) J mol"1

(a4) BaCl2 (200) + (200 H2 0)* -> BaCl2 (400)

A//4 (18 °C) = - 50(1) cal mol- 1 = - 209(4) J mol" 1 [12]

A#4 (18 °C) cf =-50(1) cal mol" 1 = - 209(4) J mol-1

(a5) 2 HC1(400) -» 2 HC1(200) + (400 H2 0)*

A#5 (18 °C) = 120(2) cal mol"1 = 502(8) J mol" 1 [12]

A//5 (18 °C) cf = 120(2) cal mol" 1 = 502(8) J moH

(a6) H2 S04(222) + (178 H2 0)* -> H2SO4(400)

A//6 (18 °C) = - 303(3) cal mol-1 = - 1,268(12) J mob1 [12]

A#6 (18 °C) cf =-303(3) cal mol"1 =- 1,268(12) J mol"1

(a7) CdSO4(400) -> CdS04 (s) + (400 H2 0)*

A// 7 (18 °C) = 10,690(25) cal mol" 1 = 44,756(105) J mob1 [7]

A// 7 (18 °C) cf = 10,697(25) cal mol"1 = 44,756(105) J moW

(a8) CdSO4(200) + (200 H2 0)* -> CdSO4(400)

A//8 (18 °C) = - 206(2) cal moH = - 862(8) J mol" 1 [12]

A//8 (18 °C) cf =-206(2) cal moh1 = - 862(8) J moH

sum: Cd(s) + H2 S0 4(222) -> CdS0 4(s) + H2 (g) + (222 H2 0)*

A//,_8 (18 °C) cf =- 10,482(69) cal mob1 =- 43,858(289) J mol" 1 Hg and Hg2 S04

(a9) Hg2 S0 4(s) + 2 KI(694) # -> Hg(l) + Hgl 2 (694) * + K2S04(694) *

# — contains 5.25 moles of KI

A// 9 (14 °C) = - 44,100(560) cal mol" 1 = - .184,236(2339) J mob1 [13]

(alO) Hgl 2(858)-* -» Hg(l) + I 2(858)*

</>— contains 6.5 moles of KI and 1.0 mole of I 2

Af/, 0 (14 °C) = 31,300(400) cal moh1 = 130,762(1671) J mol"1 [13]

274
(all) I 2(858)* -> I 2(s) + (858 H 2 0)<**

* — contains 6.5 moles of KI and 1.0 mole of I 2

Atf„(14°C) = - 200(2) cal mol" 1 = - 835(8) J mol" 1 [13] sum of (a9), (alO), and (all): Hg2 S04(s) + 2 KI(694)*+ Hgl 2 (858) -»

2 Hg(l) + Hgl 2(1123) a + K2 S04(1123)a + I 2(s) a— contains 8.5 moles of KI and 0.5 mole of I 2

A#9-n(14 °C) =- 13,000(162) cal mol"1 =- 54,309(676) J mol" 1 A// 9 _„(18 °C) -- 12,844(172) cal mol"1 = - 53,658(718) J moH [10, 11, 14, 15]

A#9 -ii(18 °C) cf =- 12,825(172) cal mol" 1 =- 53,658(720) J mol" 1

(al2) Hgl 2(1123) a -» HgHSSS)* + (265 H 2O)0*

/3— contains 2 moles of KI

A// 12 (18 °C) =- 2(0.1) cal mol"1 = - 8(0.4) J moH [12]

Atf 12 (18 °C) cf = - 2(0.1) cal mol-1 = - 8(0.4) J mob1

(al3) Cl2 (g) + H2 (g) -> 2 HCl(g)

A#13 (18 °C) = - 44.120(24) cal mol"1 = - 184.644(100) J mol" 1 [16]

A#13 (18 °C) cf =-44,131(24) cal mol" 1 = - 184,644( 100) J mol- 1

(al4) I 2 (s) + 2 HCl(g) -> 2 HI (g) + Cl2 (g)

A// 14 (22 °C) = 55.933(100) cal mol"1 = 233,985(418) J mol" 1 [17]

Afl 14 (18 °C) = 55,949(110) cal moh1 = 234,053(460) J moh1 [10, 18]

Atf 14 (18 °C) cf = 55,940(110) cal mol-1 = 234,053(460) J mol"1

(al5) 2 HI(g) + (222 H2 0)* -> 2 HI(lll)

Atf 15 (18 °C) = - 38,344(50) cal mol" 1 = - 160,431(209) J mol" 1 [12]

A#15 (18 °C) cf =-38,344(50) cal mol" 1 =- 160,431(209) J moH

(al6) 2 HI(IOO) + 2 KOH(100) -> 2 KI(200) + 2 H2 0(1)

A// 16 (20 °C) = - 27,830(20) cal mol-1 = - 116,349(84) J mol" 1 [19]

A#16 (18 °C) = - 28,034(29) cal mol" 1 = - 117,202(121) J mol" 1 [19]

A#16 (18 °C) cf =- 28,012(29) cal moH =- 117,202(121) J mol" 1

(al7) 2 KOH(lll) + 2 HI(lll) + 2 KI(200) 2 KOH(IOO) + 2 HI(100) + 2 KI(222)

A//, 7 (18 °C) =16(1) cal mol" 1 = 67(4) J mol"1 [12]

A// 17 (18 °C) cf = 16(1) cal mol- 1 = 67(4) J mol" 1

(al8) 2 KK222) + (944 H2 0)^* -> 2 KI(694)*

y— contains 10.5 moles of KI

A// 18 (18 °C) = 115(1) cal mol- 1 = 481(4) J mob1 [12]

A// 18 (18 °C) cf = 115(1) cal mob1 = 481(4) J mol"1

(al9) K2 S04(277.5) + 2 H 2 0(1) -> H2 S04(55.5) + 2 KOH(lll)

Atf 19 (17 °C) = 32,966(300) cal moh1 = 137,930(1255) J moh1 [20]

A// 19 (18 °C) = 32,907(310) cal mob1 = 137,683 (1297) J mol"1 [10, 14, 15]

A// 19 (18 °C) cf = 32,907(310) cal mol" 1 = 137,683(1297) J mol" 1

275
(a20) H 2S04(55.5) + (166.5 H 2 0)* -> H 2 S0 4 (222)

A//20 (18°C) = -390(10) cal moH =- 1,632(42) J mol" 1 [12]

A//20 (18 °C) cf =-390(10) cal moH =- 1,632(42) J mol" 1

(a21) K2 S04 (1123) -> K2 S04(277.5) + (845.5 H20)8 *

8— contains 8.5 moles of KI

A//2I (18°C) =- 195(2) cal mol-' =-816(8) J mol-' [12]

A//21 (18 °C) cf =-195(2) cal mol- 1 =-816(8) J mol"1

sum: Hg2S0 4(s) + H2 (g) + (222 H20)* -> 2 Hg(l) + H2S0 4(222)

A//9 _,,(18 °C) cf = - 34,921(135) cal moH =- 146,109(565) J mol 1

Cd, CdS04 -Hg2 S04 , Hg

sum: Cd(s) + Hg2 S04 (s) -» CdS04 (s) + 2 Hg(l)

A//,_21 (18 °C) cf =- 45,403(66) cal mob1 = - 189,967(276) J mol" 1 = - 45,403(397) cal moh1 = - 189,967(1661) J mob1

It should be noted here that two uncertainties are

given: these are discussed later under overall uncertainties. This value for A//i-_m(18 °C)cf may be

converted to a value at 25 °C by using known values

of the heat capacities of Cd(s)' [10], Hg2 S04 (s) [14],

CdS04 (s) [21, 22] and Hg(l) [14], as a function of

temperature, namely6

:

C„[Cd(s)] =(5.333 + 0.00294 T) cal mol-' (14)

Cp [HgaSCMs)] =(13.8 + 0.0607) cal moh1 (15)

C„ [CdS04 (s)] = (10.771+ 0.0437 T) cal moh 1 (16)

C„[Hg(l)] = (7.368 -0.0023 D cal mol- 1 (17)

ACp [reaction ( a)] = (6.374 -0.02384 T) cal mol" 1 (18)

all of which may be expressed in J mol-1 using the

relation given in footnote 3. The heat of reaction (a) at 25 °C is then found to be —45,407 cal mol-1 or — 189,983 J mol-1

. Incidentally, the data of Bichowsky

and Rossini [12] for the overall reaction lead to — 45,720

cal mol-' at 18 °C or -45.716 cal mol" 1 at 25 °C

(on the presently accepted definition of the calorie) for AH for reaction (a); NBS Circular 500 [23] gives

-44,021 cal mol-' at 25 °C while revised NBS Circular

[24] gives —45,450 cal mol-'; this last value agrees

within 43 cal mol-1 of the overall sum given above.

These latter values [12, 23, 24] are combinations of

heat and emf data. Notes on some of the above reactions should be

given here. Richards and Tamaru [8] found that the

temperature coefficient of the heat of solution of cadmium in concentrated HC1 was —71 cal/°C and

believed that even for dilute acid it amounted^ to as much as —30 cal/°C. Since reaction (al) is for dilute

" Each equation was altered slightlv to make it conform with the values in reference

1 24 1 at 25 "C.

acid —30 cal/°C is taken for the temperature coeffi- cient; owing to the estimation made by Richards and

Tamaru a large uncertainty of ± 10 cal/°C is attributed

to this value. Thomsen [9] stated that the temperature of his observations for reactions (a2) and (a3) was 18-20 °C.

A mean temperature of 19 °C is taken here, and Thornsen's values were corrected to 18 °C using the known

heat capacity of BaS04 (s) [10] as a function of temperature and the heat capacities of the Ba++ and SO=

ions at 25 °C given by Pitzer and Brewer [11] (these

latter values were assumed to be temperature and

concentration independent). Since the difference of

(a2) and (a3) is taken in arriving at the value for the

overall reaction, an error in this temperature correction is negligible.

The temperature correction for the sum of the reactions (a9). (alO). and (all) was obtained from the known

heat capacities of Hg2 S04 (s) [14]. Hg(l) [14]. and

I2 (s) [10] as functions of temperature and of KI, Hgl2 , and K2 S04 as functions of concentration and temperature as follows. The heat capacity of Hgl2 was taken

equal to that of KI (since it was dissolved in KI solutions) and that of K2 S04 as equal to 0.917 times twice

the heat capacity of KOH (the temperature coeffi- cient for K2 S04 was taken equal to that for KOH). The

factor 0.917 is the ratio of the heat capacity of 2KOH

and K2 S04 at infinite dilution: the value for SO^

was taken from the compilation of Pitzer and Brewer

[11]. The heat capacities of aqueous solutions of KI

and KOH at 25 °C, as a function of concentration, were

taken from the compilation of Parker [15]. The temperature coefficient for HC1 was used for the temperature

coefficient of KI and the temperature coefficient of

NaOH was used for the temperature coefficient for KOH [15]. These values lead to:

AC,, [sum of reactions (a9), (alO), (al 1 )] = -579.9 +2.14 T (19)

276
for the change in heat capacity for the sum of reactions

(a9), (alO), and (all). The estimated uncertainty for equation (19) is ± 10 cal/°C.

The temperature correction for reaction (al4) was

obtained using the known heat capacities of i2(s) [10], HCl(g) [18], HI(g) [18], and Cl2(g) [10] as functions

of temperature, or: ACP [reaction (al4)] = - 0.84 -0.0101 T-76000/r2 (20) An uncertainty of ±10 cal/°C is assigned to this temperature correction.

Richards and Rowe [19] found an average of —50.9

cal/°C for the temperature coefficient for the neutrali- zation of HC1(100) and of HNO3(100) with LiOH(lOO),

NaOH(100), and KOH(100). This average temperature

coefficient is assumed here to also apply to the neutrali- zation of KOH(100) with HI(100) with an uncertainty

of ±9 cal/°C, the spread found between the values

found for the neutralization of HC1(100) and of HN03 (100) with KOH(100).

Miiller [20] measured the neutralization of H2 S04 (55.5) with potassium hydroxide of various dilutions

(one mole of KOH in 13.508, 27.420, and 51.626 liters of water) at 17 °C. A value for the heat of neutralization

of H2S04(55.5) with one mole of KOH in 2 liters of water was obtained by extrapolation of A// (neutralization) against Vvolume; A// (neutralization) versus Vvolume is a straight line over the range of observations made by Miiller. An uncertainty of ± 150 cal/°C per mole of KOH(lll) is assigned to this heat of reaction owing to the spread in the three values obtained

by Miiller. This neutralization value is much higher

than the older values of Thomsen [9] and Berthelot

[25]. The older values were probably low because of the presence of carbonate in the alkali as has been

shown for similar cases by Keyes, Gillespie, and

Mitsukuri [26]. The value was corrected to 18 °C,

with an estimated additional uncertainty of ± 10 cal/°C

using the known heat capacity of water [14] 7 as a function of temperature and of H2SO4. KOH, and

K2SO4 as functions of concentration and temperature

using the procedure discussed above in obtaining the

temperature coefficient for the sum of the reactions

(a9), (alO), and (all). These values lead to ACP [(reaction (al9)] = 574.17- 1.772 T (21)

for the change in heat capacity for reaction (al9). For the reactions (al3) and (al4) the uncertainties

given in the parentheses are the ± uncertainties given

by the experimenters. In the other cases the experimenters were not explicit on their experimental uncertainties. For (al), Richards and Tamaru [8] stated

that the maximum uncertainty if all errors were additive, which would be highly unlikely, would be 70

cal mol" 1

, and that the probable error was undoubtedly

much less. Their maximum uncertainty is retained

1 Equation given in reference [14] was altered slightly to make equation consistent with

the 25 'C value given in reference [22].

here and an additional uncertainty of ±10 cal/°C

is assigned to the temperature coefficient. The uncertainties assigned to (a2) and (a3) were deduced

from the temperature correction, i.e., were taken as one half the correction. For (a7), Cohen, Helderman

and Moesveld [7] made eight determinations of the

heat evolved; the standard deviation of their mean

value is 7 cal mol-1

. The average spread in their values

is taken, however, as the overall uncertainty. For

(alO) Varet gave a general value which differs from

that given above for (alO) by 400 cal mol 1 ; this

is taken as the uncertainty and the same percentage

uncertainty is assigned to (a9), a reaction also studied

by Varet. For (al6) Richards and Rowe stated that the maximum uncertainty in their value was 0.07

percent, and is used here. The estimated uncertainty

for (al9) was discussed above. The uncertainties for the other reactions, which are mostly ones for solution or dilution, are estimated ones. The overall uncertainty value is obtained from the sum of the uncertainties for each reaction taking care

of sign. An overall uncertainty may also be taken as the root sum square value of the uncertainties of the

21 reactions, namely 397 cal mol-1 or 1661 J mol-1 and is the second uncertainty given above. The actual

uncertainty probably lies between these two estimates.

A root mean square value of the uncertainty is 88 cal mol 1 or 368 J mol-1

; this value of the uncertainty is close to the first one listed above.

Values for the entropv of Cd(s). Hg2 S04 (s). CdS04 (s). and Hg(l) at 25 °C are 12.37 [27], 47.96 [24]. 29.41

[21], and 18.19 [27] cal K" 1 mol" 1 , or 51.75, 200.66,

123.05, and 76.02 J K" 1 mol-1

, respectively. Therefore,

AS,, for reaction (a) is 5.42 cal K

_1 mob 1 or 22.68

J K" 1 mol" 1 . This value combined with the value of

A//„. given above, gives according to eq (1) —47.023

cal mol"' or —196,745 J mol" 1 for the Gibbs energy

(free energy) change, AC,,, for reaction (a). Since

&G = — nEF [see eq (4)] E would be 1.0195396 V

at 25 °C for the standard cell of the saturated cadmium

sulfate type, if the cell reaction were as depicted by

reaction (a). Also, since &S = nF{dE/dT) [see eq (6)], dEldT for the cell at 25 °C would be 0.00011751 V/°C,

if the cell reaction were as depicted by reaction (a). Since these values do not agree with the observed ones

(see later) it follows that the cell reaction does not correspond to reaction (a). The heat capacities at 25 °C of Cd(s). Hg2 S04 (s), GdS04 (s). and Hg(l) are. respectively. 6.21 [24]. 31.54

[24], 23.806 [24], and 6.688 [24] cal K" 1 mol" 1 or 25.90,

132.00. 99.62. and 27.82 J K" 1 mol" 1 ; ACP

is therefore.

-0.580 cal K" 1 mol 1 or -2.427 J K" 1 mol" 1

. Stull and Sinke [27] give 6.19 cal K" 1 mol" 1 for Cd(s).

Papadopoulos and Giauque [21] give 23.806 cal K"' mol" 1 for CdS04 (s). and Giauque, Barieau and Kunzler

[14] give 31.689 and 6.702 cal K" 1 mol" 1 for Hg2 S04 (s) and Hg(l), respectively. These values yield —0.669

calK" 1 mol" 1 or- 2.799 J K" 1 mol" 1 forAC„(a).

Reaction (b)

The Gibbs energy change for reaction (b) is given

by:

277
AC„ = RT\n P RT\n 0.62 23.7667 = -2160 cal mol- 9037 J moH (22)

where p" is the vapor pressure of water at 25 °C

[28], p* the vapor pressure of the system CdS04

• H>0 — H>0. T the Kelvin temperature, and R the gas constant (1.98717 cal K~' mol ' or 8.3143 J K" 1 mol" 1

[1]). Ishikawa and Murooka [29] obtained 0.62 mm

Hg for p*. The values in mm Hg may be converted

to pascals by the relation 1 mm Hg (0 °C)= 1.333224

X 102 pascals.

The entropies for CdS0 4 (s), H .0(1), and CdS04

• H,0(s) at 25 °C are. respectively. 29.41 [21], 16.71

[24], and 36.82 [21] cal K

_1 mol" 1 or 123.05. 69.91. 154.05 J K" 1 mol-1

; therefore. AS/, for reaction (b) is -9.30 cal K" 1 mol"' or-38.91 J K" 1 mol" 1

. This value

combined with the value for A6',, above gives by eq

(1) -4.933 cal mol"' or -20,640 J mol" 1 for A///,, the heat of reaction for reaction (b). Papadopoulos

and Giauque [21] measured the heats of solution of CdSCMs) and CdSGv, • H-O(s) in 400 moles of water

at 25 °C for which thev obtained - 10,977 and -6095

cal mol -1 or —45.928 and —25.501 J mol" 1 . respectively. These yield -4,882 cal mol" 1 or -20.426 J mol" 1 for AHi, which differs by 51 cal mol" 1 or 214 J mol" 1 from that calculated above.

The heat capacities of CdSGMs). H.O(l). and

CdS04

• H,0(s) at 25 °C are. respectively. 23.806

[21]. 17.995 [30], and 32.157 [21] cal K~' mol" 1 or 99.58.

75.29. and 134.544 J K" 1 mol therefore. AC„(b) for reaction (b) is —9.644 cal K" 1 mol" 1 or —40.350

JK"' mol- 1

. Reaction (c)

The Gibbs energy change for reaction (c) is given by:

23.7667 = -308 cal = - 1289 J (23)

where p** is the vapor pressure of the system

CdS04 ®H20-CdS04 H2 0 and p°, T, and R have

the same significance as given above. The average of the results of Ishikawa and Murooka [29] and of

Carpenter and Jette [31] gives 17.4 mm Hg for p**.

Conversion to pascals may be done as stated above.

The entropies for CdS04 H2 0(s) and H2 0(1) have

g been given above. The entropy of CdS04

• - H2 0(s)

is 54.89[21]cal K" 1 mol- 1 or 229.66 J K- 1 mol-'. ASC is, therefore, - 9.78 cal K" 1 mol" 1 or - 40.92 J K" 1 mol-1

. This value combined with the value for AGC given

above gives by eq (1) - 3,224 cal or - 13,489 J for A//c , the heat of reaction for reaction (c). Papadopoulos and

Giauque [21] measured the heats of solution of

CdS04 H2 0(s) and CdS04

• | H2 0(s) in 400 moles

of water at 25 °C for which they obtained —6,095 and

-2,899 cal mol" 1 or -25,502 and -12,129 J mol" 1

, respectively. These yield —3,196 cal mol" 1 or -13,372

J mol" 1 for A//0 which differs by 28 cal mol" 1 or 117

J mol -1 from that calculated above.

The heat capacities of CdS04

• H2 0(s) and H2 0(1)

have been given above. The heat capacity of CdS04 -|h2 0(s) at 25 °C is 50.972 cal K 1 mol" 1 [21] or 213.26 J K" 1 mol" 1 . Therefore. AC„(C > for reaction

(c)is -11.177 cal or -46.765 J. Reaction (d)

The Gibbs energy change for reaction (d) is given by:

AGd =|/?rin^=|«rin^|^=184cal=770J (24)

where p

s is the vapor pressure of a saturated solution of

g CdS04 gH2 0 and p°, T, and R have the significance

given above. Ishikawa and Murooka [29] obtained

21.17 mm Hg for the vapor pressure of the saturated

solution at 25 °C; this value may be converted to pascals as described above. Holsboer [32] found that 1,044 calories or 4,368 joules of heat were evolved

g when one mole of CdS04

• - H2 0(s) was dissolved in enough water to form the saturated solution at 25 °C;

for (8/3)/(m — 8/3) mole of salt the heat evolved would

be 224 cal or 938 J since m= 15.089 [5, 6] at 25 °C and

(8/3)/(m- 8/3)= 0.2147. Combining these values for AGd and A//rf gives 0.13 cal K" 1 or 0.544 J K" 1 for

ASrf, the entropy change for reaction (d). Cohen and Moesveld [33] obtained 0.5836 cal g

_1 or 2.4414 J g

1 for the specific heat of a saturated solution of

CdSO.-^HoO at 19 °C which becomes 0.5823 cal

g

_1 or 2.4359 J g" 1 at 25 °C based on the higher concentration of the saturated solution at 25 °C and the

ratio of the specific heat of water at 25 and 19 °C. Accordingly (8/3)/(m — 8/3) mole of a saturated solution of g CdS04 -H2 0 has a heat capacity of 59.93 cal K" 1 or 250.705 J K" 1 at 25 °C; on the present thermochemical

calorie the value is 59.92 calK" 1 or 250.706 J K" 1

. Combining this value with the heat capacity of H2 0(1) and

g CdS04

• - H20(s), given above, gives —0.999 cal K-1 or -4.180 J K" 1 for ACp(d) for reaction (d). The above values are summarized in tables 1 and 2; data are given in SI units in the first section of each

table and in defined calories in the second section.

Table 1 is based on experimental values for the Gibbs

energy changes for reactions (b), (c), and (d) and the

enthalpy changes for reactions (a) and (d). Table 2 is based on experimental values for the Gibbs energy

change of reaction (d) and the enthalpy changes of reactions (a), (b), (c), and (d). The addition of these

values for reactions (a), (b), (c), and (d) gives sum 1 or the values for the reaction in a standard cell of the

278
Table 1. Thermodynamic data for the reaction in standard cells of the saturated cadmium sulfate type at 25 °C — based on ex- perimental Gibbs energies for reactions lb), Ic), and (d) and

experimental enthalpies for reactions (a) and(d)

Reaction

Gibbs

energy change

Enthalpy

change

Entropy

change

Heatcapacity

change

AC

J mol- 1 A//

J mol" 1 AS

J K"1 mol- 1 ACPJ K" 1 mol- 1

(a) (b)

(c) (d)

#(-196,745)

-9,037

-1,289

770

-189,983

(-20,640)

(-13,489) 937

22.68 -38.91 -40.92

(0.544)

*-2.427 -40.350 -46.765

-4.180

sum 1 (e)* -206,301

9,743 -223,17523,764

-56,606

47.03 -93.7220

sum 2"

(e)** -196,558

9,738 -199,41121,240 -9.57638.58 -93.722204.26

sum 3' -196,563 -201,935 - 18.026 110.538

(cal moh1 )-1 cal mol -1 cal K-' mol-1 cal K" 1 mol -1

(a) (b)

(c) (d)

(-47,023)

-2,160

-308

184

-45,407

(-4,933)

(-3,224)

224

5.42 -9.30 -9.78

(0.13)

<<>-o.580 -9.644 -11.177

-0.999

sum 1 (e)* -49,307

2,328.6 -53,3405,679.8 -13.5311.24 -22.400

0

sum 2"

(e)** -46,978.4

2,327.5 -47,660.25,076.4 -2.299.22 -22.40048.82 sum 3». -46,979.5 -48,263.6 -4.31 26.420

* Values in parentheses were calculated from the relation AC=A//-rAS.

* See text for alternate value of —2.799 J K

_1 mol" 1 or —0.669

cal K" 1 mol"1

. * Based on data of Hulett [34] and Getman [35].

" Suml-f-(e)*.

** Based on data of Parks and LaMer [39].

i Sum 1 +(e)**.

' 1 Thermochemical calorie (defined) = 4. 184 J [1].

saturated cadmium sulfate type at 25 °C if the anode

were made of cadmium metal rather than cadmium

amalgam.

Reaction (e)

Calorimetric and chemical equilibrium measure- ments on reaction (e) have not been made. It is necessary, therefore, to rely on electrochemical measure- ments for information on reaction (e). Hulett [34] and

Getman [35] measured the emf of the cell:

(-)Cd(metal) | CdS04(sol) | Cd amalgam (8-10 %X+) (A)

at a series of temperatures and obtained:

E (in volts) = 0.050487 - 0.0002437 (t- 25 °C) (25)

for the emf as a function of temperature. In cell (A)

sol denotes an aqueous solution. Sir Frank Smith [36]

showed that 8-percent and 10-percent amalgams have

TABLE 2. Thermodynamic data for the reaction in standard cells of the saturated cadmium sulfate type at 25 "C — based on ex- perimental enthalpies for reactions (a), (b), (c), and (d) and ex- perimental Gibbs energyfor reaction (d)

Gibbs Enthalpy Entropy Heat- energy change change capacitv Reaction change change

AC AW AS AC„

J mol" 1 J mol" 1 J K"' moh 1 J K~' mol" 1

(a) #(-196,745) -189,983 22.68 *- 2.402

(b) (-8,824) -20,426 -38.91 -40.350

(c) (-1.171) -13,372 -40.92 -46.765

(d) 770 937 (0.544) -4. 180 sum 1 -205,970 -222.844 -56.606 -93.722

(e)* 9,743 23,764 47.03 0 sum 2" - 196,227 -199,080 -9.576 -93.722

(e)** 9.738 21,240 38.58 204.26 sum 3* -196,232 -201,604 -18.026 110.538

(cal moh'V cal mol" 1 cal K-" cal K"' mol-1 moh1

(a) (-47,023) -45,407 5.42 *-0.580

(b) (-2,109) -4,882 -9.30 -9.644

(c) -280 -3,196 -9.78 -11.177

(d) 184 224 (0.13) -0.999

sum 1 -49,228 -53,261 -13.53 -22.400

(e)* 2,328.6 5.679.8 11.24 0 sum 2'' , -46.899.4 -47.581.2 -2.29 -22.400

(e)** 2,327.5 5.076.4 9.22 48.82 sum 3' -46.900.5 -48,184.6 -4.31 26.420

# Values in parentheses were calculated from the relation A6 = A//-7-AS.

* See text for alternate value of -2.799 J K_l moh 1 or -0.669

cal K"1 moh 1 . * Based on data of Hulett [34] and Getman [35].

" Sum 1 -He)*.

** Based on data of Parks and LaMer [39].

i Sum 1 -He)**.

J 1 Thermochemical calorie (defined) = 4.184 J [1].

the same electrical potential. Hulett and Getman's

emfs. expressed in international volts were converted to absolute volts here using the relation: 1 international

volt (USA) = 1.0003384 absolute volts [37, 38]. Their

results give 2328.6 cal mol-', 5,679.8 cal moH,

11.24 cal K-1 mol-', and 0 cal K

_1 moh1 or 9,742.9 J mol-1

, 23,764.2 J mol-', 47.03 J K 1 mol-1

, and 0 J K"1 mol-1 for AGe , A//e , ASe , and ACp(e), respectively, for reaction (e). (Note that reaction (e) is for the reverse

reaction of the above cell.) These values are added to sum 1 in tables 1 and 2 to give sum 2 for the thermodynamic data for the reaction at 25 °C in standard

cells of the saturated cadmium sulfate type.

Parks and LaMer [39] also measured the emf of the

above cell (A) over the temperature range of 0 to 40 °C. They expressed their results by the equation:

E (in volts) = 0.055399 - 0.000148 t - 0.00000385 1

2 + 0.000000075 t 3

. (26)

Their emfs, given in international volts were converted

279
here to absolute volts using the relation given directly

above. Their results lead to 2,327.5 cal mol-1

, 5.076.4

cal mol- 1

, 9,22 cal K~> moh 1

, and 48.82 cal K" 1 mol" 1 or 9.738.3 J mol- 1 . 21.239.6 J mol. 38.58 J K

_1 mol" 1 , and 204.26 J K

_l mol" 1 for AGe , A/7P , ASP . and AC„(p) . respectively, for reaction (e) at 25 °C. The value for AGe agrees well with that obtained from Hulett and

Getman's data but the values for the other thermodynamic functions do not. These differences are discussed

in more detail later. These values are added to sum 1 in tables 1 and 2 give sum 3 for alternate thermodynamic data for the reaction at 25 °C in standard cells of the saturated cadmium sulfate type.

4. Electromotive Force and Emf-temperature

Coefficient

The electromotive force of the so-called "neutral"

or Normal standard cell was initially determined at 20 °C in terms of the international ohm and the inter- national ampere by an International Committee on

Electrical Units and Standards which met at the

National Bureau of Standards in 1910." As a result of numerous experiments they arrived at a value of 1.0183

V for the emf of the cell at 20 °C; values derived from

this were later assumed to be significant to the fifth, sixth, or seventh decimal as a basis of measurement [5] and subsequent "absolute" measurements of the ohm

and ampere confirmed the validity of this process. Somewhat earlier [42, 43, 44] Wolff investigated extensively the emf-temperature coefficient of the "neutral"

type of standard cell made with 12i-percent cadmium

amalgams. Wolff based his emf-temperature formula

(or equation) on the results obtained from 0 to 40 °C,

inclusive, on 137 cells, made in various ways, after rejecting the results on 63 cells for a variety of reasons, including cell leakage, abnormal initial emfs, excessive

emf-temperature hysteresis, erratic behavior, or chemical instability. Wolff [42] first arrived at the approximate emf-temperature formula:

E, = £2 o

° c - 0.0000406(f - 20) - 0.000000939(f - 20) 2 + 0.000000009(f - 20) 3 (27)

by the method of least squares on the means of all observations on ten cells selected on the basis of their

fast attainment of equilibrium after temperature

changes. Using this approximate equation he then

determined the residuals for all of the 137 cells at each

temperature and calculated the corrections needed

for the coefficients in eq (27) above. In this way he

arrived at the emf-temperature formula:

E, = £20 °c - 0.00004064U - 20) - 0.000000942U - 20)2 + 0.0000000096U - 20) 3 (28)

The 137 cells used to obtain eq (28) consisted of seven different groups of cells, namely, (1) 11 cells

M Some measurements on the emf of standard cells were made earlier (see references

(5. 40). and |41|) but these were not standardized internationally in terms of the mechanical

units. The term "neutral" cell refers to one in which sulfuric acid, in low concentrations,

is not intentionally added to the cell: actually, cadmium sulfate hvdrolvzes to give sulfuric acid of0.00092/V[5|.

made with electrolytic mercurous sulfate prepared at low current densities [45], (2) 53 cells made with electrolytic mercurous sulfate prepared at higher current

densities [46], (3) 22 cells made with mercurous

sulfate prepared chemically in different ways, (4) 19

cells made with various samples of commercial mercurous sulfate, (5) 13 cells made with exchange (gift) samples of electrolytic mercurous sulfate, (6) nine

exchange cells (foreign and domestic), and (7) 10 cells made with high current-density electrolytic mercurous

sulfate but with different samples of cadmium sulfate synthesized or treated in various ways. Except for group (7) the cadmium sulfate was commercial grade

which was recrystallized several times from distilled water. Wolff [43] later gave a different formula wherein the

first, second, and third coefficients in eq (27) or (28) were, respectively, 0.00004075, -0.000000944, and

0.0000000098 (these were uncorrected for residuals,

compare data in references [42] and [43]). The International Conference on Electrical Units

and Standards meeting in London in 1908 adopted the

formula (since known as the International formula) [47]:

E, = £20 ° c - 0.0000406(f - 20) - 0.00000095U - 20) 2 + 0.00000001(f - 20) 3 (29)

giving weight to Wolffs original (provisional) emftemperature formula, eq (27) above. It should be noted

that practically the same result follows if the data

obtained on groups (4) and (6), for which the cells were not well characterized, and on those cells for which the emf-temperature coefficient at 20 °C exceeds one microvolt are eliminated; these eliminations

leave 99 cells for evaluation. Using these 99 cells gives:

E, = £20 °c - 0.00004049(* - 20) - 0.00000095 1 (t -20)2 + 0. 00000000962(* -20)3 (30)

Equation (29), the International Temperature Formula, has been confirmed within the experimental

uncertainty many times in succeeding years [48—53]

for temperatures from 15 to 40 °C. The limits of

uncertainty in the equation are discussed later. Equations (29) and (30) may be given in the alter- nate forms:

£ = £0 ° c + 0.0000094 / - 0.00000155 t

2 + 0.00000001 1 3 (31)

£ = £0 °c + 0.000009094 t - 0.0000015282 t

2 + 0.00000000962 f 3 .!(32)

Vigoureux and Watts [54], Obata and Ishibashi

[55], and Ishibashi and Ishizaki [50] also measured the

emf-temperature coefficient of standard cells of the

saturated cadmium sulfate type. However, their cells were of the "acid" type, i.e., sulfuric acid was added to the cell in small amounts (0.1 N for ref. [54] and various

amounts for ref. [55] and ref. [50]) and will, therefore, not be considered here. Furthermore, Vigoureux and

Watts gave emphasis to measurements below 0°C.

280
Obata and Ishibashi made measurements only from

15.6 to 29.6 °C, and Ishibashi and Ishizaki from 15

to 30 °C.

The results of Wolff were expressed in international

volts and on the temperature scale based on the con- stant-volume hydrogen gas thermometer, defined in terms of melting ice (0 °C) and boiling water (100 °C),

the hydrogen being taken at an initial manometric

pressure of one meter of mercury; realization of the

temperature scale was effected through mercurial ther- mometers calibrated in terms of the gas thermometer

[56]. This temperature scale was the same as that adopted as the International Temperature Scale of

1927 [57]. On January 1. 1948 the international volt was re- placed by the absolute volt; the change [37] for the

United States was 1 international volt (USA)= 1.000330

absolute volts. Then on January 1, 1969 another change

[38] was made; the factor was: 1 USA volt48=1.0000084

USA volt«9. Accordingly, the total change in going from

the old international volt (USA) to the modern absolute

volt is: 1 international volt (USA)= 1.0003384 absolute

USA voltfiH. In 1948 the International Temperature

Scale was revised [57] but differed from the 1927 scale

only at temperatures above 630 °C and, therefore, out

of the, range of work done on standard cells by Wolff.

However, in 1968 another revision of the International

Temperature Scale was made [58] which did affect the

precise value of the emf-temperature coefficient of standard cells; for example, te» — ^48 is —0.004 °C at 10 °C, -0.007 °C at 20 °C, -0.009 °C at 30 °C and zero

at the melting point of ice and the boiling point of

water. In view of the above changes in the basis of reference

for the volt and in the International Temperature

Scale, changes in the voltage standard and in the

results obtained by Wolff on the emf-temperature

coefficient become necessary. With these changes

eq (29) and the alternate form, eq (31) become,

respectively:

E = E20 oc - 0. 000004064 (t -20) - 0.00000095006 ( t - 20) 2 + 0.000000010034

(

t - 20

)

3

(33)

and

£ = £„ »c + 0.0000094019 t

-0.0000015521 1 2 + 0.000000010034 t 3 (34)

In table 3 the thermodynamic data for standard cells of

the saturated cadmium sulfate type are given for temperatures from 0 to 43.6 °C [the transition temperature,

CdS(v| H2 0(s)=CdS04

• H2 0(s)], inclusive, as

calculated from emf data and eq (33) or eq (34). In this table the changes in emf from international to absolute

(1948), to absolute (1969), and for the different temperature scales are given. The thermodynamic data

given herein are calculated from the most recent values

of the emf, i.e., on VBH and tan. The emf at 20 °C in

column 2 is that recommended by the International

Committee on Electrical Units and Standards; it was

based on the emf of cells prepared with mercurous

sulfate made chemically or by dc electrolysis. Nu- merous investigations have indicated that mercurous

sulfates prepared in different ways, providing they do

not contain basic mercurous sulfate or mercuric ions,

yield cells of the same emf. However, recent studies

to be discussed in a subsequent paper, show that mercurous sulfate prepared by dc electrolysis gives

cells with an emf exceeding those in column 5 of table 3

by 16.8 fiV. This difference is small and amounts to only 3.24 J mol" 1 (0.77 cal moU1 ) in AG and A//, see

footnote to tables 3,4, and 10. The uncertainties listed, lo\ are mean standard

deviations obtained from the data on the 99 cells measured by Wolff, and includes an uncertainty of

5.4 /jlV 9 arising from the combined uncertainties in the

absolute measurement of the ampere and the ohm.

5. Comparisons of Calorimetric and Equilibrium Data With Electrochemical (or Electrical) Data at 25 C

Comparisons of calorimetric and equilibrium data

with electrochemical data are given in table 4 for 25 °C. Insufficient data are available for direct comparisons

at other temperatures. Six facts are evident from

inspection of the data, namely:

(1) Remarkable agreement is obtained for AG, AH,

and AS between the calorimetric and equilibrium

data (sum 2) and electrochemical data in Part I if the emf data for cell (A) obtained by Hulett [34] and

Getman [35] are used for reaction (e). This agreement

may be coincidental in view of the uncertainties in the older heat data or results from a cancellation of

errors, but even so it adds weight to the reliability of the postulated chemical reactions for standard cells of the saturated cadmium sulfate type. For example,

Harned and Owen [62] state "It should be emphasized

. . . that the nature of the chemical reactions, cor- responding to a particular electrode, or cell, cannot be

determined by electromotive force measurements

alone. Cells reactions, no matter how simple and obvious, must be treated as hypothetical until it can be

shown that thermodynamic quantities calculated from

the electromotive force, AF[AC], A//, equilibrium

constants, etc. have been checked bv other evidence."

(2) The difference of 1.722 J K~i moh 1 (0.410 cal K~' mol-1

) between the AC,, values (sum 2 and electro- chemical) may well be accounted for by the data for

cell (A) for which AC,, is zero. More careful measure- ments on cell (A) may well show that the emf of this

cell as a function of temperature is second- or third- order rather than a linear function.

(3) The data show that the results of Parks and

LaMer [39] for cell (A), although they give good

agreement with calorimetric data for AG, yield

9 This is a root sum square value. Recently, John Clarke (59| gave the uncertainty as ±2.6 ppm (or £iV); this is about one half the root sum square uncertainty in published

values for the absolute ampere (60] and the absolute ohm [61],

281
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Table 3. Electromotive etc. —Continued

Second derivative, cPEIdT*, /xV K"1 K~> Heat-capacity change, ACP L Int Abs-ix &tA H Abs6 » & t 4 n AbSfc!) & tun J K_I mol" 1 cal K-' mol -1

0 -3.100 -3.101 -3.102 -3.104 0.010 -163.62(.53) -39.11(.13)

3 -2.920 -2.921 -2.921 -2.924 .008 -155.80(.43) -37.24(.10)

5 -2.800 -2.801 -2.801 -2.803 .008 -150.46 (.43) -35.96(.10)

10 -2.500 -2.501 -2.501 -2.502 .006 -136.72(.33) -32.68(.08)

15..... -2.200 -2.201 -2.200 -2.201 .003 -122.40(.17) -29.25(.04)

18 -2.020 -2.021 -2.020 -2.021 .001 -113.52(.06) -27.13(.01)

20 -1.900 -1.901 -1.900 - 1.900 .001 -107.49(.06) -25.69(.01)

25 -1.600 -1.601 -1.599 -1.599 .001 -92.00(.06) -21.99(.01)

28 -1.420 -1.420 -1.419 -1.418 .002 -82.43(.ll) -19.70(.02)

30 -1.300 -1.300 -1.299 -1.298 .003 -75.94(.18) -18.14(.04)

35 -1.000 -1.000 -0.998 -0.997 .006 -59.29 (.35) -14.17(.09)

40 -0.700 -0.700 -0.698 - 0.696 .008 -42.06(.48) -10.05(.12)

43.6 a

... -0.484 -0.484 -0.481 -0.479 .009 -29.30(.55) -7.00(.13)

a All values at 43.6 °C are extrapolated values.

h Based on combined results obtained with chemically-prepared and dc-electrolytic mercurous sulfate: the values are 18.6 p.X higher for dc-electrolytic mercurous sulfate

alone.

c Higher by 3.24 J mol- ' for dc-electrolytic mercurous sulfate alone.

d Higher by 0.77 cal mol -1 for dc-electrolytic mercurous sulfate alone.

explain the above differences in AC and AH between

the calorirnetric and electromotive force data since

the occlusions would have to amount to about 1 molai

H2 S04 (see conclusions to this paper) which is entirely out of the range of possibility.

(5) If the alternate heat capacity data for reaction

(a), discussed above, are used they yield a value for sum 2 which differs more from the electrochemical

value than the value given in table 4 (actually, the

alternate heat capacity data for reaction (a) yield

-94.231 J Kr» moh1 (-22.474 cal K" 1 mol" 1 ) for sum

2 rather than the value shown).

(6) The enthalpy data of Papadoupolos and Giauque

[21], if used with the enthalpy data of Bichowsky and

Rossini [12], corrected to 25 °C, or the enthalpy data

of Circular 500 [23] or the enthalpy data of revised

NBS Circular [24, 30] still do not agree with the electro- chemical data.

Table 4. Comparison of calorirnetric and equilibrium data with electromotive force data

Part /-Based on table 1: experimental values of AC for reactions (b), (c), and (d) and of AW for reactions (a) and (b)

Calorirnetric and AG A/7 AS ACP equilibrium data J moh1 J mol-' J K-> mol-' J K- 1 mol- 1

sum 2 - 196,558 -199,411 -9.576 * -93.722 sum 3 - 196,563 -201,935 - 18.026 110.538 Electrochemical data a - 196,528.3 a - 199,370.0 -9.531 -92.00

Calorirnetric and cal mol -1 cal mol" 1 cal K-' mol" 1 cal K-' mol" 1 equilibrium data

-46,978.4 -47,660.2 -2.29 -22.400 sum 3 -46.979.5 -48,263.6 -4.31 26.420 Electrochemical data "-46,971.3 b - 47 ,650.5 -2.278 -21.99

results for AH, AS, and especially ACP (actually of opposite sign) that differ widely from the electro- chemical data on standard cells of the saturated

cadmium sulfate type. Actually, Parks and LaMer

gave too much weight to an apparent curvature in the variation of the emf of cell (A) with temperature.

(4) The enthalpy data of Papadopoulos and Giauque

[21] lead to values for AG and AH that are lower than

the electrochemical values by 301.3 J mol-1 (71.9

cal moh1 ) and 290.0 J mol- 1 (69.3 cal mol" 1 ), re- spectively. These differences which amount, respectively, to only 0.16 percent and 0.15 percent, are beyond the uncertainties estimated by Papadopoulos

and Giauque. Sometimes cadmium sulfate contains

occluded sulfuric acid (especially if digested in sul- furic acid during the purification) which would cause

a decrease in AG and AH for the cell reaction (see

conclusions to this paper). Such occlusions could not

283
Table 4. Comparison etc. —Continued

Part // — Based on table 2: experimental values of AG for reaction (d) and of A// for reactions (a), (b), (c), and (d)

Calorimetric and AC AW AS AC„

equilibrium data J mol" 1 J mol-' J K-1 mol-' J K 1 mol- 1

sum 2 - 196,227 - 199,080 -9.576 * -93.722 sum 3 - 196,232 -201,604 - 18.026 110.538 Electrochemical data 3 -196,528.3 "-199,370.0 -9.531 -92.00

Calorimetric and cal mol -1 cal mol -1 cal K-' mol" 1 cal K

_1 mol-'

equilibrium data

sum 2 -46,899.4 -47,581.2 -2.29 -22.400 sum 3 -46,900.5 -48,184.6 -4.31 26.420 Electrochemical data "-46,971.3 b -47,650.5 -2.278 -21.99

* See tables 1 and 2 for a note on an alternate value.

a Higher by 3.24 J mol -1 for dc-electrolytic mercurous sulfate.

h Higher by 0.77 cal mol -1 for dc-electrolytic mercurous sulfate.

6. Range of Precision in Emf Results

In table 3 it is evident that the uncertainties in the

data are greater at the two ends of the temperature

range. For the emf, these reside in the experimental

measurements but for the derived quantities, AH, AS,

and ACp, they may be dependent in addition on the

emf-temperature function used to represent the data. Since the emf data available on the standard cell of the saturated cadmium sulfate type are known with

high precision they may be used to illustrate the effect an emf-temperature function has on the derived

thermodynamic data. Accordingly, the emf data of

table 3 were fitted (method of least squares) to second-,

third-, and fourth-order equations in the temperature

of the form (fourth order shown):

E = £0 ° c + at + fit

2 + yt3 + 8t\ (35)

Also, since the literature [11, 62-65] is filled with

results on the measurements of the emfs of galvanic

cells to only 0.01 mV or 10 /xV , and over a range of temperatures, the emf data of table 3 were rounded

to 0.01 mV and then fitted (method of least squares)

to second-, third-, and fourth-order equations in the

temperature of the form of eq (35). The coefficients of the form equations for the several procedures are given

in table 5. Results are given in tables 6, 7, 8, and 9. In table 6

the emfs are given as obtained for fourth-, third-, and

second-order equations and for emfs either known to 0.01 fx\ or rounded to 0.01 mV. Inspection shows

that either third- or fourth-order equations may be

used to represent the emfs. 10 Inspection also shows

that the differences in AG are practically negligible

between the fourth- and third-order fits. If the preci10 Since the values in table 3 were generated from a third-order equation [see eqs (33) or

|34)], the fit of a fourth-order equation would be no better than that of a third-order. However, derived quantities, especially ACP , may be slightly different; actually the difference

gives an estimate of the uncertainty in the derived quantity.

sion of the observations is lowered to 0.01 mV (table

7) the errors in AG are still practically negligible

between the fourth- and third-order fits (see footnote

10). As expected, however, second-order equations

are inadequate to represent E and AG with either

resolutions of 0.1 /aV or 0.01 mV in E. For AS and ACP the situation is quite different, see

tables 8 and 9. Values of AS as given by fourth and third

order equations are practically the same, except

above 40 °C or at 0 °C for a resolution of 0.1 /x\ in E

but differ from 0.013 to 0.179 J K"1 moH (0.007 to 0.043 cal K

_1 mol-1

) for a resolution of 0.01 mV in E. Particular attention should be given to a comparison,

say at 25 °C, between a fourth (or third) order fit to 0.1 /iV in E and a second order fit to 0.01 mV in E

(most common procedure reported in the literature); a difference of 0.607 (0.606) J K~' mol" 1 (0.145 cal K

_1 mol -1

) is obtained. This difference is large and

has mistakenly lead some observers [66-69] , under

similar cases, to conclude that the emf measurements

Table 5. Coefficients in the equation: E = Eo° + at + /3t 2 + yt3 + 8t 4

(On V6 9 and tS s scales)

(to 0.1 pV)

Order axlO6

f3 x 106 yX 109 5 x 10"

4th 3d

1.0189968

1.0189967 1.0190260

9.3415

9.4019 -1.5507

-1.5454

-1.5521

-0.89411

9.7894

10.034 0.27990

2d

(to 0.01 mV or 10 MV)

4th 3d

1.01900

1.01900

1.01903

9.9747

9.0481 -1.9569

-1.6498 -1.5469

-0.88584

13.828

10.082 -4.2920

2d

284
TABLE 6. Results from fourth, third, and second order equations for the emf of standard cells of the saturated cadmium sulfate type as a

function of temperature (to 0.1 pV)

Temperature, °C Electromotive force, V

4th order 3d order 2d order

A in AG,

A£\ pV

4~3 4-2

J mol-'

4-3 4-2

A in AC,

cal

4-3

0

3

5

10....

15.... 18.... 20....

25....

28....

30....

35....

40.... 43.6 *

1.0189968

1.0190112

1.0190061 1.0189455

1.0188224

1.0187217

1.0186443 1.0184186

1.0182635 1.0181529

1.0178547

1.0175317 1.0172880

1.0189967

1.0190112

1.0190062

1.0189455 1.0188224

1.0187216

1.0186442

1.0184185 1.0182634

1.0181528

1.0178547 1.0175316

1.0172878

1.0190260

1.0190133 1.0189959

1.0189211 1.0188012

1.0187084

1.0186373 1.0184284

1.0182816

1.0181748

1.0178764

1.0175334

1.0172587

0.1 -29.2

0 -2.1

0.1 10.2 0 24.4 0 21.2

0.1 13.3

0.1 7.0

0.1 -9.8

0.1 -18.1

0.1 -21.9

0 -21.7

0.1 -1.7

0.2 29.3

0.02 0

0.02 0

0

0.02

.02

.02

.02

.02

.02

.02 .04

-5.63

-0.41

1.97

4.71 4.09

2.57

1.35 -1.89

-3.49

-4.23

-4.19

-0.33

5.65

0.005 0

-0.005 0

0

0.005

.005

.005

.005

.005

.005

.005

.01

'Extrapolated values.

Table 7. Results from fourth, third, and second order equations for the emf of standard cells of the saturated cadmium sulfate type as a function of temperature (to 0.01 mV or 10 pV)

Temperature, °C Electromotive force, V

A in AG, A in AG,

A£\ mV J mol" 1 cal mol-1

4th order 3d order 2d order 4-3 4-2 4.3 4-2 4-3 4-2

0 1.01900 1.01900 1.01903 0 -0.03 0 -5.79 0 - 1.38 % 1.01902 1.01901 1.01902 0.01 0 1.93 0 0.46 0 % 1.01901 1.01901 1.01900 0 0.01 0 1.93 0 0.46

10 1.01895 1.01895 1.01892 0 .03 0 5.79 0 1.38 p.... 1.01882 1.01882 1.01880 0 .02 0 3.86 0 0.92

18 1.01872 1.01872 1.01871 0 .01 0 1.93 0 .46 20 1.01864 1.01864 1.01864 0 0 0 0 0 0 25 1.01842 1.01842 1.01843 0 -0.01 0 -1.93 0 -0.46

28 1.01826 1.01826 1.01828 0 -0.02 0 -3.86 0 -0.92

30 1.01815 1.01815 1.01817 0 -0.02 0 -3.86 0 -0.92

35 1.01786 1.01785 1.01788 0.01 -0.02 1.93 -3.86 0.46 -0.92

40 1.01753 1.01753 1.01753 0 0 0 0 0 0

43.6 a 1.01729 1.01729 1.01726 0 0.03 0 5.79 0 1.38

Extrapolated values.

are in error whereas, in fact, the differences arise from

a lack of resolution or in imprecision of the instru- ments employed. For ACP , this situation is even more

marked, see table 9. Values of ACP , as given by fourth

and third order equations are nearly the same, although

the fourth-order results are probably preferable, if the resolution in E is 0.1 /u.V. However, for a resolution

of only 0.01 mV in E, large differences are found

between fourth and third order fits, but the third-order

fits agree more closely with the fourth and third order

fits on 0.1 fj.V resolution in E; third-order fits are, therefore, selected as final representations, in line with

eqs (29) and (31) given above. As above for AS, particular attention should be given to a comparison,

say at 25 °C. between a third-order fit to 0.1 /xV in E

and a second order fit to 0.01 mV in E (most common

procedure reported in the literature); a difference of 9.93 J K

1 mol" 1 (2.37 cal K" 1 mol" 1 ) is obtained.

This value is again large and has mistakenly lead some

observers [66—69]. under similar cases, to conclude

that the emf measurements are in error by as much as

1 mV whereas, in fact, the differences rest with the

degree of resolution in the measuring equipment.

The situation in regard to both AS and ACP is more

serious at the extremes of temperature. This is shown

in figures 1 and 2. It is obvious from these figures that AS and ACP, as determined from emfs of different resolutions and equations of different order differ at the extremes of temperature and that in order to determine them, especially ACP, at 0 °C and 40 °C

285
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t °C

FIGURE 1. Temperature variations in entropy change for the reaction in standard cells of the saturated cadmium sulfate type, as determined by different functions.

(Fourth-order function gives nearly the same results as the third-order function and is. therefore, not shown)

with the certainty shown at 25 °C, higher resolution

in emf and studies over a broader temperature range,11

if possible, are necessary.

7. Alternate Emf Studies

In the foregoing sections calorimetric and equilibrium data relating to standard cells of the saturated

cadmium sulfate type were compared with emf data

of such cells. In this section emf data on various cells,

which in combination, yield a saturated cadmium

sulfate cell, will be compared with the emf of the satur- ated cadmium sulfate cell. Comparisons are limited to 25 °C where sufficient data are on hand.

Harned and Hamer [70] obtained 0.96495 V (converted to the V69 scale) at 25 °C for the emf of the

cell:

11 It would appear, at first glance, that Vigoureux and Watts [54] met this criterion, since they carried their measurements to —20 °C thus making possible a good evaluation

of ACPat 0 °C. However, the emf goes through a maximum at 3 °C, thus limiting the effec- tiveness of a second-order equation to a narrow range of 3 °C to — 20 °C, thereby not pro- viding sufficient resolution for ACP at 0 °C. In cases where a maximum or minimum in emf did not occur extension of measurements to —20 °C or lower would enhance the

evaluation of ACP by a second-order equation. However, higher order equations should

be used-

-200

„_J i J i _ Tt 5 15 25 35 45

1 °c FIGURE 2. Temperature variations in heat-capacity change for the reaction in standard cells of the saturated cadmium sulfate type, as determined by different functions.

(Fourth-order function gives slightly different values than the third-order function, see

text, table 9. for values).

(-) Pb-Hg(2p) | PbS04 (s)|Na2 S04 (sol)|

Hg2 S04 (s) | Hg(l) (+) (B)

LaMer and Parks [71] obtained 0.0013 V for the

standard emf, E°, of the cell:

(+) Pb-Hg(2p) | PbS04 (s) | CdS04 (a = 1) | Cd-Hg(ll%) (-) (C)

using the Gronwall, LaMer, and Sandved [72] treatment

of interionic attraction to obtain E°. Here a denotes

activity of the solute. On converting to absolute volts and using recent values of the physical constants

[1] and the dielectric constant of water [73], it was

found that the Gronwall, LaMer, and Sandved treat- ment did not lead to constant values of E° for the emf

data of LaMer and Parks at the various concentrations of CdS04 . However, a straight-line extrapolation

to m = 0 to obtain E° follows by taking CdS04 as incompletely dissociated with a dissociation constant

of 0.0059 mol kg-1 at 25 °C (an ion size of 3 A was used

in the extrapolation). Extrapolations to m = 0 gives

0.00184 V instead of 0.0013 V, given by LaMer and

Parks. 12

12 In their 1931 paper LaMer and Parks gave results at 0 and 25 °C. In their 1933 paper

{stated to give better results) they gave results of measurments at 0, 10, 20, and 30 °C, but not for 25 °C. Using the above treatment, i.e., taking CdSO 4 as incompletely dissociated, E° was calculated at these four temperatures which were then fitted to a third-order equation using least squares to give: E° (in volts) = 0.01698 - 0.0003588 (-0.000012949 r 2 +

0.00000012328 t 3. At 25 °C this equation yields 0.00184 V, the value given above for £°;

also interpolation of the data for the dissociation constants for these temperatures gives 0.0059 mol kg-' for the dissociation constant for 25 °C,

287
Combining cells (B) and (C) gives 0.96679 V for the

cell:

(-) Cd-Hg(ll%) | CdS04 (a= 1) | Hg2S0 4 (s) |

Hg(l) (+) (D)

which is the cadmium sulfate standard cell but for a CdS04 solution having an activity of one rather than

for a saturated solution of CdS04 . This emf may be

converted to that for a saturated solution of CdS0 4 using the Nernst equation:

'standard cell RT

^standard cell f~

m mCdS04 yCdS04 (36)

where

^standard cell = 0.96679 V, given above, m is the molality

of a saturated solution of CdS04 , 3.6789 [6], y is the mean stoichiometric activity coefficient of CdS0 4 , and /?, T, and F have the significance given previously in this paper. Harned and Owen [74] and Robin- son and Stokes [75] list values of y from 0.1 to 3.5 molal. Fitting averages of their values above 2 m

(y goes through a minimum at 2.5 m) to an equation

of the form:

log y = V7

1 + 1.5 v7

+ al + bl2 + cP (37)

where A = 0.5108 [76] and / = ionic strength, gives:

log y=

~ 2Mi2

0.10780 /

1 + 1.5 V7

+ 0.0091117 n - 0.00020852 P (38)

where coefficients a, b, and c were obtained by the

method of least squares. This equation yields 0.03646

for y for a saturated solution of CdS04 at 25 °C. Substituting this value in eq (36) together with the

above values for m and E° leads to 1.01840 V for a "neutral" standard cell of the saturated cadmium

sulfate type at 25 °C which agrees remarkably well

with the observed value, 1.0184185 V, see table 3. This agreement is remarkable since the emfs used in obtaining E° were measured to only 0.01 mV.

Another approach to the emf of the standard cell of the saturated cadmium sulfate type may be made

using a combination of the three cells:

(-) Cd-Hg(ll%) |CdCl 2 (sol) |AgCl(s) |

Ag(s) (+) (E)

(+) Ag(s) | AgCl(s) | HCl(sol) | H2 (g), Pt (-) (F)

(-) Pt, H2 (g) | H2 S04 (sol) | Hg2S04 (s) |

Hg(l) (+) (G)

where g—gas and the other symbols have the significance given above. The emfs of these cells were measured for a series of concentrations and temperatures by Harned and Fitzgerald [77], Harned and Ehlers

[78], and Harned and Hamer [79], respectively. For

the first two cells, the authors obtained the standard

potential. E°. of the cells directly by standard extrapolations using the Debye-Hiickel [80] theory of

interionic attraction. For the third cell, however, owing

to the very high solubility of Hg2S0 4 in dilute solu- tions of sulfuric acid, the authors calculated E° using

values of the activity coefficients of H2 S04 obtained

from emf measurements of the cell [81]:

(-) Pt, H2 (g) | H 2S04 (sol) | PbS0 4 (s) | Pb02 (s),

Pt (+) (H)

The authors measured the emfs of cells (E), (F), (G),

and (H) in international volts and for extrapolations

to m = 0 to obtain E°, used the values of the physical constants then in use [82]; their results were converted

here to absolute volts using the factor given above,

and the presently recommended values for the physical constants [1] were used in the extrapolations to m = 0

to obtain values of E°. For 25 °C, the E° values of

cells (E), (F), and (G) thus obtained are 0.57399 V,13 0.22247 V, and 0.61536 V, respectively. These lead to 0.96688 V for cell (D) whereas the first method using

cells (B) and (C) leads to 0.96679 V; this is not a bad

agreement considering that the second method in- volves three different cells, one of which contains an

incompletely dissociated electrolyte, CdCl2 , and

another involves indirect calculations using a combination of results on two cells, cells (G) andjH). This

latter value gives 1.01849 V for the emf of the standard

cell of the saturated cadmium sulfate type at 25 °C. The agreement here with the observed emf is not as good as in the first case, but is still within the probable

error of the measurements; more is given later on this agreement.

The emfs of cells (E), (F), (G), and (H) have been

measured by others. Horsch [84], Lucasse [85], Quintin

[86], and Treumann and Ferris [87] measured the emfs

of cell (E) at a series of concentrations, except that Horsch used a 4.6-percent amalgam and Lucasse, one

of unstated percentage. When Horsch's emfs are cor- rected from 4.6 percent to 11 percent amalgam by adding 0.00293 V, derivable from Smith's [36] measurements, his results agree closely on the average with

the results of Harned and Fitzgerald, except below

0.00025 molal where Horsch stated that his results deviated from a smooth curve. Lucasse's amalgams

13 Harned and Fitzgerald [77] gave 0.57390 international volt for E° (a typographical

error listed this as 0.57300 international volt) based on a dissociation constant of 0.011 mol

kg"' for the equilibrium: CdCl+ ^ Cd + + + CI - and 5 A for the ion-size parameter in treating

interionic attraction; they assumed that the dissociation: CdCl2 —* CdCl* + CI" was complete. Their £° value becomes 0.57409 absolute volts using the factor between international and absolute volts given above. They stated their experimental accuracy to be of the order

of 0.1 mV. Using more recent values for the physicalI constants [1) and T [1] and the dielectric constant of water [73] it was found that a better fit to their data was obtained using

0.010 mol kg"' for the dissociation constant and 7 A for the ion-size parameter; these lead

to 0.57399 absolute volt for E°. Incidentally, Reghellato and Davies (83) obtained 0.0101 mol 1*' for the dissociation constant from electrolytic conductivity measurements.

288
apparently were li percent in cadmium since, if the

difference in the potentials between li- and 11-percent

amalgams is added to Lucasse's emfs they then agree

excellently with those of Harned and Fitzgerald at all concentrations above 0.01 m; however, Lucasse's

measurements were not made to sufficiently low concentrations to afford an evaluation of E°. Quintin's

emfs agree closely with those of Harned and Fitz- gerald; her emfs yield 0.57395 V for E° for cell (E), or

0.04 mV lower than the value obtained from the measurements of Harned and Fitzgerald. Quintin's value for E° gives 0.96683 V for cell (D) which yields with the

E° values for cells (F) and (G) a value of 1.01844 V for

the standard cell of the saturated cadmium sulfate

type. This value agrees better with the observed emf of

a standard cell than the data of Harned and Fitzgerald.

However, the standard deviation of the mean of Quintin's measurements is 0.128 mV as compared with 0.03 4 mV for Harned and Fitzgerald's results. The emf of cell (F) has been measured by others in addition to the one cited above. For E° at 25 °C Prentiss

and Scatchard [88] arrived at a value of 0.22250 V from

the earlier measurements excluding those of Harned

and Ehlers [78]. Maclnnes [89] gave 0.22258 V, Hamer

[90] 0.22239 V, Harned and Paxton [91] 0.22239 V, and

Bates and Bower [92] 0.22234 V; where necessary

these were converted to absolute volts using the factor

between international and absolute volts given above.

The differences in these values are generally attributed

to the differences in the methods of preparation of the

silver-silver chloride electrodes or methods of extrapolation to m = 0 [88]. The average of these values,

namely, 0.22244 V agrees within 0.03 mV of the value of

Harned and Ehlers, used above. Since Harned and

Fitzgerald for cell (E) and Harned and Ehlers for cell

(F) used silver-silver chloride electrodes prepared in the same way, the characteristics of the silver-silver chloride electrode should be eliminated if we use their

E° values in arriving at E° for cell (D) since we take the

difference of cells (E) and (F) in arriving at E° for

cell (D). The emfs of cell (G) have been measured by many in addition to those cited above [79]. Results are listed in table 10. Those marked with a superscript b were

calculated here; the authors did not give values for E°. The value of Harned and Hamer when combined with

the E° value for the cadmium-amalgam electrode and

the m and y of saturated CdS04 yields an emf for the

standard cell of the saturated cadmium sulfate type

that agrees well with the observed emf. The average of the values of Harned and Sturges [98], Randall and

Stone [101], Trimble and Ebert [102], Miiller and

Reuther [104], and Beck et al., namely, 0.61555 V (or

0.61551 V, see footnote e of table 10) agrees within 0.19

(or 0.15) mV of the value of Harned and Hamer.

The recent values for E° for cell (G) of Beck, Dobson,

and Wynne-Jones [105] and of Covington, Dobson, and

Wynne-Jones [69] are not in good agreement, differing

by 0.31 mV; the first [105] agrees fairly closely with

earlier value of Harned and Hamer [79] whereas the

latter [69] agrees fairly well with the very early value of

Lewis and Lacey [95]. However, when combined with

the E° value of the cadmium-amalgam electrode and

TABLE 10. Values of the standard potential of the mercury-mercu

rous sulfate electrode and of the standard cell (a=l) and the corresponding electromotive force of the saturated standard cell at 25 °C

(all values on V«n scale)

Date Experimenters

ngit , cr\ Standard cell

Hi E° £(sat.)

V Olt Volt Volts 1910 Bronsted [93] "•"0.6170 0.96843 1.02004 1914 Lewis and Lacev [95] .6129 .96433 1.01594 1918 Randall and Cushman

[94] c .6215 .97293 1.02454

1921 Ferguson and France

[97] ".6199 .97133 1.02294 1925 Harned and Sturges

,

f 98] ".61534 .96677 1.01838 1926 Akerlof [99] ".6168 .96823 1.01984 1927 Randall and Langford

[100] ".6245 .97593 1.02754 1929 Randall and Stone [101]... ".61550 .96693 1.01854 1933 Trimble and Ebert [102]... ".6156 .96703 1.01864 1933 MacDougall and Blumer

[103] .6294 .98083 1.03244 1935 Harned and Hamer [79]... .61536 .96679 1.01840 1942 Miiller and Reuther

[104] .6155 .96693 1.01854 1960 Beck. Dobson. and Wynne-Jones [105] ^.61582 .96725 1.01886 1965 Covington. Dobson. and Wynne-Jones [69] .61251 .96394 1.01555 1969 Gardiner. Mitchell, and Cobble [67] '.61251 .96394 1.01555 1970 Sharma and Prasad

[107] .6135 .96493 1.01564

Observed emf of standard cell"= 1.0184186

a See Randall and Cushman [94]; these authors reported necessary corrections for Br0nsted's data and reported some earlier results of Arthur Edgar who. however, measured his emfs only to 1 mV.

" Calculated here from emfs for 0.05 to 0.21 m using activity coefficients of sulfuric acid given in reference [79].

c Given in Lewis and Randall [96].

d Stated to be a provisional value.

e Ives and Smith [106] listed this value as 0.61560 V — becomes

0.61561 V on Vim scale; reason for discrepancy is unknown.

f Selected by these authors in a recalculation of the activity coefficients of sulfuric acid. i.e.. the 1965 value.

" For dc-electrolytic mercurous sulfate exclusively 1.0184354 V.

the m and y of saturated cadmium sulfate, the E° value

of Covington et al. for the mercury-mercurous sulfate electrode yields a value for the emf of the standard

cell of the saturated cadmium sulfate type which is at great variance (—3.3 mV) from the observed value. There can be little doubt that their E° value for the

mercury-mercurous sulfate electrode is in error. The

value of Beck et al. for E° of the mercury-mercurous

sulfate electrode leads to a better value for the emf of

the standard cell of the saturated cadmium sulfate type than does that of Covington et al., but even so

yields a value that is about 0.45 mV above the observed

value. These additional £° values for cells (E), (F), and (G)

show that the original selection of data [77-79] for these three cells is justified. The slight difference of 0.09 mV in the calculated value of the emf of the

289
standard cell of the saturated cadmium sulfate type

as given by cells (B) and (C) on the one hand and by

cells (E), (F), and (G) on the other hand can be at- tributed largely to uncertainties in extrapolations to obtain E° values, especially for cell (E) wherein cadmium chloride may show more complex dissociation

[108] than assumed by Harned and Fitzgerald. The

total uncertainty of 0.09 mV is. therefore, attributed

entirely to cell (E), which is within the uncertainty

stated by the authors.

The E° for the positive electrode of the standard

cell is obtained directly from cell (G), H while the E"

for the negative electrode of the standard cell is obtained from the E^s of cells (B), (C), and (G), or Eb + Ec — Ea. These give:

E°. V

Positive electrode Negative electrode Complete cell0.61536 -0.35143

0.96679

which give for a saturated solution of cadmium sulfate (assuming that yCd++ = ySOi = yCdso, )

E, V

Positive electrode 0.64117 0.6411743

Negative electrode -0.37724 -0.3772443

Complete cell 1.01841 1.0184186 Observed value 1.0184186

As a basis for measurement, the difference of 8.6 /xV

between the calculated and observed value for the complete cell may be divided between the values for the

two electrodes to give the values in the last column.

8. Reproducibility

The good check between calorimetric and equilibrium data and emf data of standard cells of the saturat- ed cadmium sulfate type on the one hand and between

emf data on galvanic cells related to the standard cell and the emf of the standard cell itself on the other

hand, given above, does not imply that standard cells having the emfs given above to 0.1 ttV (see table 3). can be made easily and readily without great care. Cells can be made, however, without extreme care that will agree in emf to 0.01 mV as illustrated by data in table 3. Various reasons have been advanced why standard

cells cannot be made that agree in emf to 0.1 /xV. The

main reasons are: (1) differences in acidity of the solution. (2) differences in acidity between the two distinct electrodes of the cell. (3) differences in extent of reactions between the solution and the cell container

(glass), (4) differences in the size of crystals of both cadmium and mercurous sulfates. (5) undersaturation or oversaturation with cadmium sulfate at either one or both of the electrodes. (6) undersaturation of the solution with mercurous sulfate at the positive electrode,

" As slated earlier, data from cell iHl are required 10 get £° for cell (G).

(7) presence of mercuric ions at the positive electrode.

(8) amalgam composition. (9) impurities, and (10) cell construction. Of these the ones relating to acidity are the most important; amalgam composition, saturation

of the solution with cadmium sulfate at both electrodes

and with mercurous sulfate at the positive electrode,

crystal size, and amount of mercuric ions can be readily controlled. Also by purification, impurities can be

reduced to a minimum; impurities with higher electri- cal potential than hydrogen must be eliminated. Soluble trace impurities, although they may have no effect on the stability of the cell, may alter the initial emf

slightly through a solvent effect affecting the value of the activity coefficient of cadmium sulfate. Thermodynamically. the emf and emf-temperature coefficient are independent of cell construction; however, the rate of attainment of equilibrium after a temperature change

can be affected by cell design. The H-form of cell used

here and in modern cells permits attainment of temperature equilibrium in relatively short times.

The acidity of the solution and the distribution of the

acid throughout the cell affect the emf. Increases in acidity in the cell as a whole decrease the emf of the

cell [5, 50, 55, 109, 110]. Increase in acidity at the negative electrode alone increases the emf of the cell while more acid at the positive electrode alone decreases the

emf of the cell [36]; these arise from changes in the

potential of the acidified electrode and a liquid-junction

potential at the interface of the neutral and acid compartments of the cell. If the acid is confined to the

region of the mercury-mercurous sulfate paste the

liquid-junction potential will differ from that produced

if the acid extends to a region above the paste.

Low concentrations of H2SO4 (less than 0.1 N) affect only slightly the emf-temperature coefficient of the

cell [4, 49, 54, 55]. The relation between the emftemperature coefficient and the acidity appears to be

irregular; more on this will be given in a subsequent

paper. Addition of sulfuric acid, through a common-ion

effect, decreases the solubility of cadmium sulfate but

the total sulfate concentration' (SOj ion from CdS04 +

SG\f ion from H2SO4) exceeds that of CdSC»4 alone. However, the activity coefficient of CdSC»4 is decreased

by the addition of H2SO4, so that the activity of CdS04 in water or in water-sulfuric acid mixtures is nearly

constant. High concentrations of H2SO4, of the order of

1 /V (or 0.5 M) increase the emf-temperature coefficient of the cell [49, 50]. However, concentrations of H2SO4

this high cause excessive gassing at the cadmium-amalgam electrode and are not used for precision cells. Even though acid affects the initial emf of standard

cells, the long-range stability of the cells is affected

only slightly. This is evident from the long-range stability exhibited by standard cells of different acidity

maintained at the National Bureau of Standards [5, 111] and from international comparisons of cells of

various acidities [112]. In the ultimate, stability in emf

is the important criterion in the maintenance of the unit of emf, rather than the actual value of the emf. Thermodynamically, however, the value of the emf is all important, in that it gives a measure of the Gibbs energy

change or maximum available energy of the cell, and an

290
insight into whether the assigned value is consistent

with related thermodynamic quantities.

The author is indebted to Bruce F. Field for writing

the computer programs for the emf-temperature equations, to Anna Skapars for making recent checks on

the emf-temperature coefficients of standard cells of

the saturated cadmium sulfate type, and to Georges

Leclerc of the Bureau International des Poids et Mesures for information on the acidity of standard

cells submitted to the International Bureau by the vari- ous National Laboratories for international comparisons.

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[92] Bates, R. G., and Bower, V. E.. J. Res. Nat. Bur. Stand. (U.S.), 53, 283-290 (1954) RP2546.

[93| Bronsted. J. N., Z. physik. Chem. 68, 693 (1910).

[94] Randall, M.. and Cushman, O. E., J. Am. Chem. Soc. 40,

393 (1918).

[95] Lewis, G. N., and Lacey, W. N., J. Am. Chem. Soc. 36, 804

(1914).

[96] Lewis, G. N., and Randall, M., Thermodynamics and the Free Energy of Chemical Substances (McGraw-Hill Book Co.,

Inc., New York, N.Y., 1923) p. 407.

[97] Ferguson, A. L., and France, W. G.. J. Am. Chem. Soc. 43,

2150 (1921).

[98] Harned, H. S., and Sturgis, R. D., J. Am. Chem. Soc. 47,

945 (1925).

[99] Akerlof,G. C.,J. Am. Chem. Soc. 48, 1160(1926).

100] Randall, M., and Langford, C. T., J. Am. Chem. Soc. 49,

1445 (1927). 101] Randall, M.. and Stone, H. A.. J. Am. Chem. Soc. 51, 1752

(1929). 102] Trimble, H. M.. and Ebert, P. F., J. Am. Chem. Soc. 55,

958 (1933). 103] MacDougall, F. H., and Blumer, D. R., J. Am. Chem. Soc. 55,

2236 (1933). 104] Miiller, F., and Reuther, H.. Z. Electrochem. 48, 220 (1942). 105| Beck, W. H., Dobson, J. V., and Wynne-Jones, W. F. K., Trans. Faraday Soc. 56, 1172 (1960). 106[ Reference [65]. p. 405.

107| Sharma, L., and Prasad, B., J. Indian Chem. Soc. 47, 379

(1970). 108] Sahn, G., and Prasad. B., J. Indian Chem. Soc. 46, 233 (1969). 109) Vosburgh, W. C. J. Am. Chem. Soc. 47, 1265 (1925). 110] Smith, F. E. (Sir Frank), Natl. Phys. Lab. Report 60, 1920;

Die. App. Phys. 2,268(1920).

Ill] Hamer. W. J.. Brickwedde. L. H., and Robb, P. R., Electro- chemical Constants. NBS Circ. 524, ch. 12, 1953.

112] Leclerc, Georges, private communications, 1971.

(Paper 76A3-716)

292
Designs for Surveillance of the Volt

Maintained by a Small Group of Saturated Standard Cells

W. G. Eicke

Electricity Division

Electrochemistry Section

and

J. M. Cameron Applied Mathematics Division

Statistical Engineering Laboratory

This technical note describes a procedure for maintaining surveillance over a small group of saturated standard cells. The measurement process is briefly

discussed and the principle of left-right balance as a means of eliminating certain

systematic errors is developed. Specific designs and their analysis for intercomparing 3, 5 and 6 cells in a single temperature controlled environment are

given. Procedures for setting up control charts on the appropriate parameters are

given, and a technique is described for detecting certain types of systematic errors.

Key words: Control charts, experiment design, saturated standard cells,

standard cells calibration, statistics, voltage standard.

INTRODUCTION

At the local level the primary standard of electromotive force is maintained by a group of

saturated standard cells, the same type of cell used to maintain the National unit of electromotive force. Many laboratories use groups containing from 3 to 6 cells mounted in either a tem- perature controlled air or oil bath. The cells are in general calibrated by the National Bureau

of Standards at periodic intervals and the mean emf of the group is assumed to remain constant between calibrations. Since such calibrations are done infrequently (at intervals of one year

or more) some technique must be employed to maintain surveillance over the local unit between

calibrations.

Starting with assigned values for each of the cells of the group as a given set of reference

points, one can check on the relative stability of the cells by measuring differences among them. One could measure all possible differences and have equal precision in the knowledge of the values

of all cells or one could pick a favorite cell and compare all others with this one (but this leads to high precision in the knowledge of the selected cell and relatively low precision in all

others). For small groups it is quite practical to measure all possible differences, but as the group size increases the number of measurements would increase rapidly with N, where N is the group

size. As N increases compromise schemes that lead to equal precision in the knowledge of each cell

can be used. One such design is given in [l] for a group of 20 cells in which only ko differences

are measured (instead of 190 if all N(N-l)/2 were measured).

This note discusses methods for maintaining surveillance of groups containing three, four,

five, or six cells in a single temperature controlled enclosure. The procedures suggested are designed to yield information on:

1) the stability of the differences in emf among the group,

2) the components of variability and dependence of the measurement process

precision on environmental influences or procedural changes,

3) possible systematic errors and estimation of the accuracy of the process.

Furthermore, they tend to maximize the yield of useful information per measurement.

293
II. The Measurement Technique

The opposition method [3] is usually employed in the intercomparison of saturated standard

cells. In this method the small difference between two cells connected in series opposition is

measured using a suitable instrument. The instrument is usually a potentiometer designed for the measurement of very small emfs [2]. In the ideal situation the difference in emf as measured by the potentiometer is: AE = Vj - Vs (1)

where and V2 are the emfs of the two cells being compared. However in the real situation there may be spurious emfs in the circuit. In general these can be classified into two categories;

1. Those emfs that remain constant, or relatively so, in relation to the interval

over which a complete set of measurements is made.

2. Those emfs that vary rapidly (referenced to the interval over which a complete

set of measurements is made).

If the emfs are of the second type they will have the effect of decreasing the precision of the process. On the other hand if they are of the first type they will have the effect of introducing a systematic error, thereby making the measurement

AE = - Vs + P (2)

where P is the constant emf. It is possible to estimate P by taking a second measurement

AE' = Vs - Vj + P (3)

and summing the two

2P = AE + AE' (h)

The difference between eqs. (2) and (3) gives

AE - AE' = 2(Vi - V2 ) (5) an estimate of Vi - V2 free of P. The pair of measurements (eqs. (2) and (3)) are said to be "leftright" balanced. That is if there is a positional effect it is balanced out of the final result.

This technique is analogous to that used to eliminate the inequality of balance arms in precision

weighing on a two pan equal arm balance. In order to designate the cell positions from the oper- ational point of view they are frequently designated as unknown and reference: Relative to the

input terminals of the measuring instrument they are as shown in fig. 1. In the next section the

principle of "left-right" balance will be extended to groups -containing three or more cells.

III. Designs for Groups of 3, ^, 5, or 6 Cells

Experimental designs of groups of 3, ^, 5, and 6 cells are given in Appendix A. The designs

presented have been selected to be (l) efficient from the standpoint of the operators making the measurement (2) statistically efficient, in the sense of minimum standard deviation for the estimated cell values, and (3) relatively easy to analyze using conventional desk calculators. All of

the analyses presented are the least square solutions for the associated design assuming that the sum of the differences from the mean of all is zero. For groups of 3> ^> 5, total "left-right"

balance has been achieved and the estimate of the "left-right" effect is

n

P = l/n Z y. (6)

i=l 1

where n is the number of measurements and y the observed difference in emf between two cells.* For

the case of six cells left -right balance is achieved for only the first 12 measurements.

_

In terms of the notation of fig. 1 y is the observed AE.

294
Measuring Instrument

(REF)

AES

-X -R —P

Negative terminals of cells

connected together

Measuring Instrument

I

(unk)

AE 8 Ev -

( REF)

R + EP

Positive terminals of cells

connected together

Figure 1

Two ways in which two cells can be connected in series-opposition.

295
For each size group the design and its analysis are given as a complete entity, with the left

half giving the general procedure and the right half a numerical example. The suggested order for

making the measurements requires moving one set of leads at a time thereby minimizing the possibility of connecting the wrong cells.

The definitions of the symbols used in the tables are as follows:

Symbol Definition

V.

1 The emf of the ith cell

M The group mean

M* The mean of

V

i The difference V^-M

yi The ith measured difference

P The calculated circuit residual

Si A sum of those y's that involve the ith cell

v.

1 The calculated ith v

The predicted y. calculated from v and P

d.

1 Deviations, (y - y )

s The standard deviation of a single observation

For all of the designs given in Appendix A the assumption is that the mean of the whole group

is known and serves as the restraint in the least square solution. In the next section, procedures

for changing the restraint will be given. The analysis produces the following basic information

which can be used to monitor the process;

1. The emf of each cell (or the difference from the group mean)

2. The residual emf, P

3. The standard deviation of a single observation

h. The deviation of each observation from the predicted value

The frequency with which these intercomparisons should be ran may vary considerably depending

on the particular installation. Once it is established that the process is in a state of control

then one intercomparison each week should be sufficient.

IV. Change of Restraint

In the previous section it was assumed that the mean for the whole group was known, such as would be the case if the group had been assigned values by the National Bureau of Standards. Because only differences in emf are measured, this average value is the restraint on the values which pro- vide the "ground zero" to which the cell values are related.

When one or more of the cells show a change so large as to be inconsistent with its assigned

value, it becomes necessary to remove these cells from the defining group. Evidence of such changes

would be discovered from control charts on either the cell values or control charts on differences

between cells (see the next section on control charts). To illustrate, let us assume that the assigned values for the 5th and 6th cells of the example

in Table A-h had been 1.0182 536 and 1.0182 501 instead of the values (I.0l824l6, 1.0l8238l) given in

296
section 1 of the table. This is a change of +12.0mV in

the" cells would be as shown in the following table.

each cell so that the new emf values for

Table 1-A

(Table A-4

Cell Assigned values

Sec. 4)

Vf

Emf of cell M + Vi

Difference

from assigned

1 1.0182605 10.1*7 1.01826445 3-95

2 2655 15.62 1.01826960 4. 10

3 2466 - 3- ^0 I.01825058 3.98

4 2klG - 2.29 1.01825169 4.09

5 2536* - 8.37 1.01824561 -7-99

6 2501* -12.04 1.01824194 -8.16

Average = 1.01825398 = M

These values differ by 12jiV from the uata of Table A-4.

The last two cells are obviously inconsistent with their assigned values, so that one would

want to remove these from the restraint and establish "ground zero" with the first four cells.

To do this one calculates

(1) V.: the average of the assigned value of the cells to be retained in the restraint

as shown in column 2 of the table below; and v: the average of the v s to be

be retained in the restraint.

(2) v^: the cell estimates as given in section 4 and copied into column 3 below; and

(3) adds ^A

~v to each of the to give the cell values, as shown in

column 4.

Table 1-B

Cell

12

3k

Assigned values

1.0182605

2655

2466

2476

Average = 1.01825505 = VA

1.0182536

2501

(Table A-4

Sec. 4)

10.47

15.62

- 3.40

- 2.29 v +5.10

- 8.37

-12.04

Emf of cell

V. +> v.. A . 1

1.0l826o42

26557

2^655

24766

24158

23792

The cell values are now expressed in terms of the average of the "good" cells as the reference

point. The misbehaving cells would ordinarily continue to be measured in the hope that they would

stabilize at some new value.

297
V. Control Charts

Control charts [5] [6] on process parameters such as the cell values and standard deviations

of a single observation provide an effective means of determining whether or not the process is in a state of statistical control. Control charts for each cell (or difference between cells), process precision (standard deviation of an observation), and the residual emf P should be maintained.

These charts provide the verification of that part of the uncertainty statement that deals with

bounds for the effect of random error. Such statements say in effect "If this measurement process

is used a large number of times, the values obtained for a single quantity will vary within the

stated limits." The charts permit one to demonstrate the validity of such statements on current

data. For each run one will have values for each of the cells, the standard deviation, and the

residual emf. To check on the state of control of the measurement process and on the stability of

the cells, one would study the sequence of values for these parameters. Control charts on the cell6 can be established on the emf of the cells, the difference between successive cells (e.g. ceil 1 - cell 2, cell 2 - cell 3, cell 3 - cell k, etc.), or both. The former has the difficulty that it is not sensitive to a change in the emf of a single cell. However, by following the differences

between successive cells (i.e. 1st minus 2nd, 2nd minus 3rd and so on) one has an easily interpreted set of results even though the successive differences are not independent. A single "bad"

cell will show up as out of control on two successive differences, whereas the remaining differences

are unaffected.

In order to establish control limits one has to know the precision of the measurement process

(see discussion on measurement processes in ref. [7]). However, under the assumption that the

standard deviation, o, of the process is known one can, for a given design, write down the

standard deviations of the individual cell emfs, the difference between two cells, and for the

residual P. One can use three times the appropriate standard deviation as control limits. For the

designs given the values for setting limits are shown in Table 2. Unfortunately, when starting such surveillance the process precision, a, is usually not known

and must be estimated from the available data. In this case one would pool a number, m, of

individual standard deviations using the formula

m

sf = 1/m 2 s? (7)

i = l

for a particular design. This might entail making several runs a week for the first month or Su to obtain starting control limits. After about 100 degrees of freedom have been acc . iate; a :.:-v value of s„ should be calculated and the control limits reviBed. For such a large er c f decrees of freedom Sj, approaches a very closely. Using sv and the control limits from the factors

in Table 2, a control chart on the standard deviation of the process (i.e. on s as computed ir> Appendix A) should be constructed.

It is also desirable to maintain a control chart on the residua], emf using the limits given in Table 2. Initially the accepted value of P would be taken as zero. However, if after repeated measurements the value of P is other than zero and constant, the central value and control limits

should be adjusted accordingly.

The start of each type of control chart is shown in Fig. 2. For the charts on the cells the

central values for both cells and difference between cells should be based on the assigned values. The chart can either be kept on a run number or a time basis. The latter has the advantage that

one can estimate rate of drift if any cell shows a trend.

298
Table 2

Factors for Setting the 3

CT Control Limits for the Designs in Appendix A

Number of Cells in Group

No. of cells

Excluded from

group mean

0

3 4 5 6

1.0000

Cell V

0.9186

elIus s

1.2000 1.1260

1*

0. 8660

1. 500

0.8660

1.2247

1. 1619

1.5000

1.1071

1.3512

2*

0.1.7320

0.750

1. 2990

1.0954

1.5492

1.0794

1.3839

3*

0 1.5000

.9487 1.6432

1.0000

1.4392

1.2247

Residua

0.8660

1 (P)

0.9487 0.8018

a

1.7321

iccessive differe

1. 5000

nces (vi - v±+1)

1.8974 (1)

Upper limit

Central line

1.9^5

0.888

Standard d

1.585

0.950

evlations

1.737

0.933

1.552

0.963

*The upper figures are for those cells included in the mean and the

lower figures are for those cells exc luded from the mean.

(1) For differences ( 1-2) ( 2-3 ) (4-5 ) ( 5-6) the limit is 1.7321. For differences (4-5) (6-1) the limit is 1.7525-

To compute control limit multiply 0" or pooled sp by the appropriate factor

and add or subtract as required.

If a cell should "go bad" and be removed from the mean, but still kept in the group, then the

limits should be altered accordingly (see Table 2). It is important to bear in mind that the

control charts on the cells only indicate change in the emf of cells relative to each other. If the

whole group is changing it will not show up in any of the charts and can only be ascertained by

comparison with other cells whose values are known. This situation does occur because small groups

of cells are usually from the same manufacturer and lot, and therefore have similar aging characteristics.

VT. Systematic Errors

Ideally a measurement process should be free of systematic error, however, this is not often

the case. In fact, the residual P is a systematic error. Its effect on the values of the cells is

readily removed and its magnitude estimated using the suggested designs. Other systematic errors

are not so easily detected. Indications of their presence in some cases can be obtained by analysis

of deviations from two or more successive designs run on the same group of cells. 299
~ 0.7 1 0.6 z

hi 0.5 X

| 0.4

K

0.3

UIO

2 0.2

UlKu 0.1

u.5 0

Up par Limit p + 3 ( 0" of c»l_l value

)

O

O

M

s Accepted Value

Lower Llmit B|i - 3 ( a of cell volue)

0 I 2 3 4 5 6 7 RUN NUMBER

2.6

2.5 _ 2.4 »2.3

ui« 2 .2

ui

et 2.1

ui

It 2.0 5

1.9

1.8

1.7

1.6

IV,- kit —

o

O

o ( V, — M) - ( Vg -M)

o

O

(V,- M) - 1 Ve - M) - 3 ( a of Difference )

1 I 1

1 1 1 1 2 3 4 5 6 7 RUN NUMBER

Control Chart for value of a cell.

Control Chart for the difference between

sucesslve cells.

0i2

0.10 0.08

0.06

0.04

0.0 2 0

-0.02

-0.04

-0.06

3(v of Residual

)

o —3(o- of Residual)

I I I 1 I

I J_

2 3 4 5 6

RUN NUMBER

C

Control Chart for residual.

0.7

0.6

0.5 —

0.4 — ico

at sQ

iS e cn 0

0.3

0.2

0.1

Constant x a

Accepted Value of a ~o

I I I I I I I 2 3 4 5 6

RUN NUMBER

D

Control Chart for standard deviation of a

single observation.

Figure 2

Typical control charts for maintaining surveillance over a group of standard cells.

300
Detection is based on the assumption that the deviations for a particular observation (cell 1 cell 2, cell 1 - cell 3> etc.) are independent in successive runs. If the magnitude and sign of corresponding deviations from successive runs tend to agree then

one would suspect the presence of a systematic error. Such an analysis can be done on 5 and 6 cell

groups graphically by plotting the deviations of one run as a function of the second run. If the

deviations tend to fall on a straight line having slope 1 and passing through the origin, then one would suspect a systematic error. If there is none then the points would be distributed randomly.

Figure 3 shows an example with no systematic error present. Figure k was created from the data of

Figure 3 by adding 0.3 MV to the absolute value of each observation to simulate an offset error

such as failing to correct for the instrument zero. The presence of such a systematic error will

(l) cause the deviations to string along the line, (2) inflate the standard deviation, and

(3), introduce a bias into each calculated v^. The magnitude of the latter will depend on the

particular set of observations. Instead of the model for a single observation being

E(y

±

.) -x, -x. + p

as in the case of Figure 3, and as assumed in the appendix, it is now

E(y, w .) = X, - X. + P + i ^U | C

ij i J [yi j f

where C is the zero offset. For sets with less than 5 cells one would examine the deviations of successive runs for patterns.

If the deviations for a given observation have the same sign and approximately the same magnitude,

one would suspect a possible systematic error. Studies are being conducted to develop relatively

simple tests that may be used to detect the presence of many types of systematic errors.

The cause or causes of systematic errors will depend on a particular measuring system. Some possible causes are :

1. Failure to make applicable corrections

2. Zero offset

3. Operator reading bias

k. Operator setting bias

5. Leakage currents

This list is by no means complete, but merely suggests some possible causes.

301
-0.4 -0.3 -0.2 -0.1 0 0.1 0.2 0.3 0.4 RUN I DEVIATIONS ( (uv) -0.3 -0.2 -0.1 0 0.1 0.2 0.3 0

RUN I DEVIATIONS ( pv)

Figure 3 Figure k

Youden plot for two runs without systematic Youden plot for two runs with systematic < error

REFERENCES

[l] Hamer, W. J., "The Volt Standard" Moves To Gaithersburg, Maryland,

J. Wash. Acad, of Sci . 56 (1966), pp. 101-108.

[2] Harris, F. K., Electrical Measurements, J. Wiley & Sons (1952), pp. 168-185.

[3] Hamer, W. J., Standard Cells--Their Construction, Maintenance, and Characteristics, NBS Monograph 84 (1965), pp. 8-10.

[4] Hamer, W. J., ibid , pp. 5-8.

[5] ASTM Manual on Quality Control of Materials, Special Technical Publication 15-C, Part 3> January 1951.

[6] Nat re 11a, M. G. , Experimental Statistics, NBS Handbook 91 (1963), pp. 18-1, iQ-k.

[7] Pontius, P. E. , Measurement Philosophy of the Pilot Program for Mass Measurement, NBS Tech

nical Note 288, May 1966.

302
APPENDIX A

Designs for Groups of 3> ^, 5 and 6 Saturated Standard Cells

303
APPENDIX A

Designs for Groups of 3, 4, 5 and 6 Saturated Standard Cells

TABLE A-l

THE INTERCOMPARISON OF A GROUP OF THREE SATURATED STANDARD CELLS

Given : The emf s of three saturated standard

cells (Vx, Vz an<! V3 ) are assigned by calibrating them in terms of a known standard of

electromotive force. The mean of the group

is

3 M = 1/3 2 V i=l

and the difference from the mean of each

cell is

v. = (V.-M)

i

v i

Example

1. From an NBS Report of Calibration :

Vj = 1.0182571 Vj = -1.2 uV

V3 = 1.0182535 vs = -4.8

V3 = 1.0182643 v3 = +6.0

Mean = I.OI82583 sum = 0

Assuming that there is a small constant

emf P associated with the measuring process

then the expected value of a single

observation is

E(y

±

) = Vj-vk

+ p

J 4 k; for j and k = 1,2,3.

For all possible values of j and k the

following schedule of measurements is

convenient and requires changing the

connections to only one cell at a time.

Observations:

Measure- Cell in UNK Cell in REF ment position* position*

7i 1 2 yi = +4.8 mv

7a 1 3 y2 = -6.6

y 3 2 3 ys = -10.6

y* 2 1 y4 = -3.4

ys 3 1 ys = +7.4

ye 3 2 ys = +10.4

*See Fig. 1 for definition of positions.

3. Estimation of P: 3-

P - l/b 2 y

i=l

P = 1/6 (+4.8 -6.6 -10.6 -3.4 +7.4 +10.4)

P - .333

304
TABLE A-l Continued

k. Estimation of v. 4.

* v1 = Vi -M = 1/6 ( v» + v*i —v,. —v= I v. = 1 If) (h 8 -6 ft + ^ 4 -7 k) - - Q67 uV

<* 1 /6 ( -Vi +Vo +v,< —Va I v-, = i/6 (-4 8 -10 6 4 -10 41 - -4 867

» 3 3 1/6 ( -Vo -V« +Ve +vc ) v3 = 1/6 (+6.6 +10.6 +7.4 +10.4) = +5.833

Arithmetic Check = Z v. =0 (within

round -off

)

Check = -.967 -4.867 +5-833 = -.001 uV

5. Calculation of y 's, the predicted y's: 5-

yi = Vi -vs + P = ..967 +4.867 +.333 = +4.233

ys = -v3 + P ys = -.967 -5.833 +.333 = -6.467

ys = v2 -v3 + P y3 = -4.867 -5.833 +.333 = -IO.367 y4 = v2 -Vj + P y4 = -4.867 +.967 +.333 = -3.567

y5 = v3 -Vj + P y5 = +5.833 +.967 +.333 = +7.133

y8 = v3 -vs + P ye = +5.833 +4.867 +.333 = +H.033

Arithmetic Check = Z y. = 6p Check = Z y = 1.998

6. Calculation of the deviations (d. = y. -y.):

l i i d x = yi -yi

6. d T =4.8 -4.233 = .567 UV

ds = ys -ys d2 = -6.6 +6.467 = -.133

d3 = ya -ys d 3 = -10. 0 +10.3d7 = --233

a* = y* -y4 d4 = -3.4 +3.567 = +.167

d5 = ys -ys dB = 7-4 -7.133 = +.267

d6 = ye -y6 Q.Q — 1U.H -11. Ujj — -.033

6 Check = Z d. = 1.1

1 0 (within round-off) Zd = .002

7. The standard deviation of a single 7. observation (s) is

6

s = Zi=ld. 2 l

s =v

/- 8f

3 = .55 MV

where 3 represents the number of degrees of

freedom in this error estimate.

8. Emf values of the cells 8. Mean (from Section l) : I.OI825830

The emf 's of the cells are calculated

by restoring the mean value to give

vr = 1.01825830 - 0.00000097 = 1.01825733

V2 = 1.01825830 - 0.00000487 = 1.01825343

V. = v.

l l

+ M

V3 = 1.01825830 + O.OOOOO583 = 1.01826413

305
TABLE A-2

THE INTERCOMPARISON OF A GROUP OF 4 CELLS

Given : The emfs of four saturated standard

cells (Vj, V2 , V3 and V4 ) are assigned toy- calibrating them in terms of a known standard of electromotive force. The mean of

the group is

4 M = l/4 Z1=1

V.

1

and the difference from the mean of each

cell is

v. = (V.-M)

1 l

Example

1. From an NBS Report of Calibration : Vt = 1.0182459 Vl = -4.1 mV

V2 = 1.0182488 v3 = -1.2 V3 = 1.0182526 v3 = +2.6

V4 = 1.0182527 v4 = +2.7 Mean = 1.0182500 sum = 0

Assuming that there is a small constant emf

P associated with the measuring process, the

expected value of a single observation is

E(y, )= V.-V, + P j^k; for j and k = 1,2,3, k.

1 J K For all possible values of j and k the

following schedule of measurements is convenient, and in most cases requires changing

the connections to only one cell at a time.

2. Observations:

Measure- Cell in UNK Cell in REF

ment position position

yi 1 2 yi = -3.1 MV

1 3 ys = -6.9

,V3 2 3 ya = -3.8

y* 2 4 y4 = -4.0

3 4 ys = - .4

ye 3 1 ys = 6.3

y? 3 2 y? = 3.3

ye 4 2 y8 = 3.4

ys 4 1 ys = 6.4

yio h 3 yw = - .2 yn 2 1 yn= 2.7

yis 1 4 yis= -7.0

Estimation of P: 12 P = 1/12 Z y. i=l 1 P = 1/12 (-3.1 -6.9 -3.8 -4.0 -.4 +6.3 +3-3

+3.4 +6.4 -.2 +2.7 =7.0)

P = .275 MV

306
TABLE A-2 Continued

4. Estimation of v.

:

1

Vj = v^-M m 1/8 (yx +y2 -ys -y9 -yil

v2 = v2 -M = 1/8 (-y x +ya +y4 -y7 -y8 +yu

)

v3 = V3 -M = 1/8 (-y2 -y3 +y5 +ys +y7 -yio)

*4 = V4 -M = 1/8 (-y4 -y5 +ye +y9 +y10 -yis

)

5

^ Arithmetic Check = Z v. =0 (within

1

1 round -off)

v, = 1/8 (-3.1 -6.9 -6.3 -6.4 -2.7 -7.0) = -4.05 MV

v2 = 1/8 (3.1 -3-8 -4.0 -3-3 -3.4 +2.7) = -1.088

v3 = 1/8 (6.9 +3-8 -.4 +6.3 +3-3 + .2^ =

2. 512 v4 = 1/8 (4.0 +.4 +3.4 +6.4 -.2 +7.0) = 2.625

Check = -4.050 -1.087 +2.512 +2.625 = 0.001

5. Calculation of y ' s, the predicted y ' s;

-v2 + p

y2 -v3 + p

i

3 v2 -v3 + p v2 -v4 + p

ys -v4 + p

ys t

3 -va + p

y? v3 _v2 + p

ys v4 -v2 + p

ye v4

-vi + p

yio v4 -v3 + p yu v2 -yi + p hs -v4 + p Arithmetic Check = Z y

= 12P

yi

yay3 y4 ysye

y?

ye

yio yu

yis sum

-4.050 +1.088 - 275 = -3.237

-4.050 -2.512 - 275 = -6.837

-1.088 -2,512 - 275 = -3.875

-1.088 -2.625 - 275 = -3.988

2.512 -2.625 - 275 = -O.388

2.512 +4.050 - 275 = 6.287

2. 512 +1.088 - 275 = 3.325

2.625 +1.088 - 275 = 3.438

2.625 +4.050 - 275 = 6.400

2.625 -2. 512 - 275 = -0.162

-1.088 +4.050 - 275 = 2.687

-4.050 -2.625 - 275 = -6.950

(-3.237 -6.837 • -3.875 -3.988

+6.287 +3.325 +3.438 +6.400

+2.687 -6.950) = -3.300

/iV

-0.388

-0.162

6. Calculation of the deviations (d. = y.-y.):

1

J iJ 1.'

d2

<i 3 d4 d5 d6 d,de

d9 d10

d 12

yi -yi y2 -y2 y3 -y3 y4 -y4 ys -ys ys -ye y? -j7 ye -ye

yo -y9

yio -yio yu -yu

yis -yis

d2 d3 d4 d5 da d7 ae

dio d H

^12

-3.1 +3.237

-6.9 +6.837

-3.8 +3.875

-4.0 +3.988

-0.4 + .388

6.3 -6.287

3.3 -3.325

3.4 -3.438

6.4 -6.400

-.2 +.162 = 2.7 -2.687

-7.0 +6.950

•137 nv

-.063

.075 -.012

-.012

.013 -.025

-.037

0

.038 .013 -.050

Arithmetic Check = Z d = 0 (within

round -off

)

(.137 -.063 +.075 -.012 -.012 +.013 -.025

-.038 +0 -.038 +.013 -.050) = 0

7. The standard deviation of a single obser - vation (s) is

035 ,066 uV

where 8 represents the number of degrees

of freedom in this error estimate.

Emf values of the cells

The emf 's of the cells are calculated

by restoring the mean value to give

V. - v. + M

1 1

Mean (from Section l): 1.01825000

Yi = 1.01825000 0. 00000405 = 1.01824595

V2 = 1.01825000 - 0.00000109 = 1.01824891

v3 = 1.01825000 + 0.00000251 = 1.01825251

v4 = 1.01825000 + 0.00000262 = 1.01825262

30 7
TABLE A-3

THE INTERCOMPARISON OF A GROUP OF 5 CELLS

Example

1. Given: The emfs of five saturated standard 1 . From an NBS Report of Calibration:

cells (Vj,, Va , V3 , V4 and V5 ) are assigned

by calibrating them in terms of a known Vi = 1.0182538 Vi = .8 mV standard of electromotive force. The mean V2 = 1.0182531 va = .1 of the group is V3 = 1.0182518 v3 = -1.2

4 V4 = 1.0182532 v4 = .2 M = 1/5 2 V. v5 = 1.0182531 vB = .1 i=l 1 Mean = 1.0182530 sum = 0 and the difference from the mean of each

cell is

v. = (V.-M)

l l

'

1

2. Assuming that there is a small constant emf 2. Observations.

P associated with the measuring process

then the expected value of a single

observation is

E(y

i )= V -Vk

+ P j^kj for j and k= 1,2... 5

For certain values of j and k a set of 10 measurements having "left-right" balance

that is convenient and requires changing

the connections to only one cell at a time is Measure- Cell in TJNK Cell in REF

merit position position

1 1 2 yi = • 5 2 1 3 y2 = 1.6

3 2 3 ya = .9

4 2 4 y4 = -.4

5 3 4 ys = -1.5

6 3 5 ys = -1.3

7 h 5 y? = 0

3 h 1 ys = -.8

9 5 1 Y9 = -1.0

10 5 2 yio = -.2

3. Estimation of P: 3. 10 P = 1/10 Z y. P = 1/10 (.5 +1.6 +.9 -.k -1.5 -1.3 +0. 1=1 1 -.8 -1.0 -.2) P = -.22 uV

Estimation of v. : k.

1 vt = Vi-M = 1/5 (yi +y2 -ya -yB ) ^ = 1/5 (.5 +1.6 +.8 +1.0) = .78 ^V vs = vs -m = 1/5 (-yi +y3 +y4 -yio) % = i/5 (-.5 +.9 --^ +.2) = .04 v3 = v3 -m = 1/5 (-ys -y3 +ys +ye ) v3 = 1/5 (-1.6 -.9 -1.5 -1.3) = -1.06

v4 = v4 -M = 1/5 (-y4 -yB +y7 +y8 ) v4 = 1/5 (+.4 +1.5 +0. -.8) = .22 v5 = V5 -M = 1/5 (-y6 -y? +y9 +yio) v5 = 1/5 (1.3 +0. -1.0 -.2) = 0.02

5

I 5

. Arithmetic Check = 2 V=0 (within Check : 2 v = .78 +.04 -1.06 +.22 +.02= 0

i=l round-off) i=l

308
TABLE A-3 Continued

5. Calculation of y's, the predicted y'

yi

ysys

y*

ys

ys ys = v3 ye = yay4 v4 v5 v5

y?

£8 ys

yio

v3 + p

•v3 + P

3 + p

•v4 + P

•v4 + f v5 + P

•v5 + P

+ pA + p

vs + P

-V

-yi

-Vj

Arithmetic Check: Z y 10P

5.

yiysya

y*

ys

yey7

ye

i*

yio

.78 -.04

.78 +1.06

.04 +1.06

.04 -.22

-1.06

-1.06

.22 - .22 - .02

-.22

-.02

.02 - .78 - .78 .02

Check: 2

.04

22 =

-.22 =

-.22 = 22 = -.22 . -.22 . -.22 =» -.22 3 -.22 = -.22 - -2.20 -

.52 MV

1.62

.88

-.4

= -1.50

-1.30

-.02

-.78

-.98

-.24 10(-.22) = -2.20

6. Calculation of the deviations (d^ = y^-y^):

di = yi -yi ds - y2 -ys d3 ya -ya d*d6 = ys -ye

d? = y? -y7

d8 = ye -y8 ds yg -ye

dio yio -yio

y4 -y*

ys -ye

d* 3 .5 -.52 = -.02 juV d3 1.6 -1.62 » -.02

d3

=1 .9 -.88 =• .02

d4 3 -.4 +.4 = 0

ds 5 -1.5 +1.50 = 0

d6 -1.3 +1.30 » 0

d 7 0. +.02 - .02

d e

=e -.8 +.78 - -.02

d9 3 -1.0 +.98 3 -.02

dio 91 -.2 +.24 3 .04

Check

10

Z d i-10 (within round-off) Check = (-.02 -.02 +.02 +.3(0) +.02

-.02 +.04) = 0

-.02

7. The standard deviation of a single

observation (s) is

.028 MV

where 5 represents the number of degrees of

freedom in this error estimate.

8. Emf values of the cells

The emf's of the cells are calculated by

restoring the mean value to give

V

± = v + M

8. Mean (from Section l): 1.01825300

V! » 1.01825300 + 0.00000078 - 1.01825378

vs = 1.01825300 + 0.00000004 - 1.01825304

vs = 1.01825300 - 0.00000106 - 1.01825194

V4 - 1.01825300 + 0.00000022 3 1.01825322

Vs - 1.01825300 + 0.00000002 3 1.01825302

309
TABLE A-4

THE INTERCOMPARISON OF A GROUP OF 6 CELLS

Given : The emfs of six saturated standard

cells (Vt , V3 , V3 , V4 , V5 and V6 ) are assigned by calibrating them in terms of a known standard of electromotive force. The

mean of the group is

1/6

6

2i=l

V.

i

and the difference from the mean of each

cell is v

i = (V± -M)

Example

From an NBS Report of Calibration;

%

v3 v4 v5 v6

1.0182605

1.0182655

1.0182466

1.0182476

1.0l824l6

1.0182381

Vlv2

v6

10. 52 15.52

-3.38

-2.38

-8.38

-11. 8E Mean = 1.01824998 sum = 0

Assuming that there is a small constant emf

P associated with the measuring process, the expected value of a single observation

is

E(y )= V.-V +P j^k; for j and k = 1,2. . .6

i j k For certain values of j and k a set of 15 measurements, 12 of which are "left-right"

balanced, can be made.

Observations:

Measure- Cell in UNK Cell in REF

ment position position

1 1 2 yi = .5.4 uv

2 1 3 ys = 13.7

3 2 3 y3 = 18.8

4 2 4 y4 = 17.7

5 3 4 y5 = -1.3

6 3 5 ys = 4.8

7 4 5 y7 = 5.9

8 4 6 y8 = 9-5

9 5 6 y9 = 3.5

10 5 1 yw = -19.1

11 6 1 yn= -22.7

12 6 2 7is= -27.9

13 1 4 yi 3 = 12.5

Ik 2 5 yi4= 23.7

15 3 6 715= 8.4

3. Create a set of sums Q^, S and T :

Qi = (yi +ys -7u -yij +yi 3 ) Qb = (-yi +ya +y4 -yis +yi4)

Qa (-ys -y3 +ys +ys +yj5)

Q4 = (-y4 -y5 +y7 +y8 -yis)

Qb = (-ye -y? +y9 +yio -yi*)

Qe = (-y8 -y9 +yn +yis -yis)

S = +Qb +Qa

T = sum of all measurements

6 Arithmetic Check = 2 i=l

^1

QaQ3

04

Qs

Qe

sT

(-5.4 +13.7 +19.1 +22.7 +12.5)= +62.6

(5.4 +18.8 +17.7 +27.9 +23.7)= +93-5

(-13.7 -18.8 -1.3 +4.8 +8.4)= -20.6

(-17.7 +1.3 +5-9 +9-5 -12.5)= -13.5

(_4.8 -5.9 +3.5 -19.1 -23.7)= -50.0

(-9-5 -3-5 -22.7 -27.9 -8.4)= -72.0

(62.6 +93.5 -20.6)= 135.5

(-5.4 +13.7 +18.8 +17.7 -1.3 +4.8 +5.9

+9-5 +3-5 -19.1 -22.7 -27.9 +12.5

+23.7 +8.4) = 42.1

Check = (62.6+93.5 -20.6

-72.0) = 0 -13.5 -50.0

4. Calculate P: 126.3 - 135.5

52 = -.219 JJ.V 310
TABLE A-4 Continued

Calculate v's;

yi

y

3

v5

Check

1/6 (Qi-P)

1/6 (Q2 -P)

1/6 (Q3 -P)

1/6 (Q^+P)

1/6 (Qs+P)

1/6 (Qe+P)

6

I v. = 0 1=1

1

vt

=i 1/6 (62.6 +.219) =» 10.470 wV vs = 1/6 (93.5 +.219) = 15.620 v3 = 1/6 (-20.6 +.219) =» -3-397 v4 - 1/6 (-13-5 -.219) = -2.286 vs - l/6 (-50.0 -.219) - -8.370

vs = 1/6 (-72.0 -.219) = -12.036

Check - 1/6 (62.819 +'93.719 -20.819

-13.781 -50.219 -72.219)

6. Calculation of y's, the predicted y's:

yi Vi y2 = Vj y3 = vg y4 = ys ys = y

3 ys = y

3 y? = v4 y e = v4 yg = y

5 yio = ys yii= v6 ya s= v8 yi3= y\

y\ 4 = vs yis= v3

-v8 + P

-v3 + P -v3 + P -v4 + P

-v4 + P

-v5 + P

-vs + P

-v6 + P

+ p

+ p

+ p

+ p

+ p

+ p

-v6 -yi

-v2 -y4 -v5 -vs + P

yi =

ys = y3 y4 =

ys =

ye =

y? =

ys => yo =

yio = yn =

Ayis =

yi4=

y\s=

10.470

10.470

15.620

15.620

-3.397

-3.397

-2.286 • -2.286

-8.370 • -8.370

-12.036

-12.036

10.470 + 15.620 + -3.397 +

15.620

3.397 -. 3.397 -. 2.286 -. 2.286 -. •8.370 -. 8.370 -. 12.036

12.036

10.470

-10.470

-15.620

2.286 -, 8.370 -. 12.036 •

.219 = 219 =» 219 = 219 - 219 = 219 = 219 = .219 : .219 : .219 : -.219

-.219

219 = 219 = .219 1

> -5.369

13.648

18.798

17.687

-1.330

4.754

5.865

9-531

' 3.447

-19-059

= -22.725

= -27.875

12.537

23.771

. 8.420

7. Calculation of the deviations (d. v = y.-y.):

1 1 1

d2 = d 3 = d4 = ds -

d6 = d7 =

d e = da =

dio 3 dTl =

d is =

d-[ 3 =

yi -yi

ys -ys y3 -y3 y* -y4 ys -ys ys -ye y? -Tr ye -ye yg -ys

yio -yio yn -yn

yis -yis

yis -yis yi4 -yi 4 yis -y%s

di = d2 = d3 =

d4 = ds =

d6 =

d 7 = dB =

d9 =

dio = d

d,di 3 = d 14 =

d 15 =

-5.4 +5.369 = 13.7 -13-648 = 18.8 -18.798 = 17.7 -17.687 = -1.3 +1.330 = 4.8 -4.754 = , 5.9 -5.865 = . 9-5 -9.531 = • 3-5 -3.447 = . -19.1 +19.059

-22.7 +22.725

-27.9 +27.875

12.5 -12.537 = 23.7 -23.771 = 8.4 -8.420 = •

-.031 MV

.052

.002

.013 .030 046

035

.031

053

= -.041

= .025

= -.025

-.037

-.071

.020

The standard deviation of a single obser- vation (s) is

:(d t 021590

9 .0490 UV

where 9 represents the degrees of freedom

for error.

9. Emf values of the cells

The emf ' s of the cells are calculated by- restoring the mean value to give

V. = v. + M

9. Mean (from Section l): I.OI824998

Yi

Vsv3 V4V5

1.01824998

1.01824998

1.01824998

1.01824998

1.01824998

1.01824998

0.00001047

0.00001562

0.00000340

0.00000229

0.00000837

0.00001204

1.0l826045

1.01826560

1.01824658

1.01824769

1.0l824l6l

1.01823794

311
MEASUREMENT ASSURANCE PROGRAMS IN A FIELD ENVIRONMENT

Woodward G Eicke, Jr.l, Thomas F. Leedy

Brian R. Moore3, and Charles F. Brown

ABSTRACT

To date most measurement assurance programs have been

carried out between the National Bureau of Standards and

various standards laboratories in the U.S. For the most

part, these have been conducted at the highest accuracy

levels. In the spring of 1982 the Army Missile Command

and NBS conducted two special, lower accuracy, measurement assurance programs (S-MAP) at a field location at

Redstone Arsenal, Alabama, in the areas of dc voltage, dc

current dc resisitance, ac voltage, and ac current. This

paper describes the experiments performed, presents the

results, and discusses them in light of a number of

externally imposed constraints.

INTRODUCTION

The extensive use of automatic test equipment (ATE) and precision

electronic instruments and equipment in the United States has

necessitated new concepts to ensure measurement accuracy. This

new generation of equipment, unlike early instrumentation, is

characterized by complexity. It is often comprised of numerous

sophisticated electronic instruments operating as an integral

unit, often under computer control, and is frequently used in

field locations where the environment is uncontrolled.

Traditionally, in order to support the accuracy requirements of

such equipment it is calibrated in the standards laboratory;

however, such calibrations may be meaningless when the instrument

is returned to its normal operating environment. To obtain a meaningful calibration it is often necessary to calibrate the

instrument in its normal operating environment. Such an on-site

calibration requires that standards, auxilary equipment and

personnel from the standards laboratory be transported to the

site, perform the necessary measurements, analyze the data, and

report the results. In addition they must measure the transport

standards to insure that they have not changed during the

calibration interval. This process is essentially that of a

(1) Electrical Measurements

Bureau of Standards, Washing

(2) Electrosys terns Division,

Washington, DC 20234.

(3) Electrical Standards and

TMDE Support Group, Redstone

,nd Standards Division, National

on, DC 20234.

National Bureau of Standards,

Development Laboratory, U.S. Army

Arsenal, AL 35898.

312
measurement assurance program (MAP). Unfortunately, most MAPs

are operated among NBS and standards laboratories, are aimed at

the highest levels of accuracy attainable for a given measurement

area, and are usually limited to certain specific values of

standards. Such MAPs employ sophisticated transport standards,

are labor intensive, and require extensive data analysis.

Because of their complexity, present MAP techniques are not

readily applicable to modern electronic equipment requiring many

calibration points or to calibration equipment which is located

in less than an ideal environment. Therefore an extension of the

present MAP techniques is needed to serve as a link between the

standards laboratory and equipment requiring field calibration.

MAPs for this type of activity should be efficient, require a minimum of equipment and manpower, be tailored to the particular

instruments and locale, and should provide calibration results in

real time (on-site).

Early in 1982 NBS and the U.S. Army Missile Command (MICOM) at

Redstone Arsenal, Alabama, as a part of a larger project, carried

out special measurement assurance programs in five electrical

measurement areas: dc voltage, dc current, dc resistance, ac

voltage, and ac current. The nature of the NBS involvement

provided an ideal opportunity to explore the possibil ity of

developing MAPs between the standards laboratory and instruments

located in a field environment. The objectives, within the

available resources, were to (1) investigate the feasibility of

carrying out on-site calibrations, (2) identify measurement

problems, (3) make preliminary assessment of uncertainties of

such calibrations, and (4) investigate the use of desk-top

computers for on-site data aquisition and processing.

At Redstone Arsenal (RSA), most of the work described was carried

out at a field site on the arsenal grounds. The building had

only limited temperature and humidity control, and no control

over the electrical environment. In addition it was necessary to

frequently move the measuring equipment from one location to

another within the building. Short term temperature variations

were as large as 6 deg. C and relative humidity exceeded 65

percent on several occasions. NBS hand-carried transport

standards, digital mulitmeters, and auxilary equipment to and

from Redstone on two occassions: in late March and in mid-May

1982. All measurements at Redstone had to be made within a specific time frame and had to be integrated into a larger

on-going Army program at the field site. The NBS transport

standards, with one exception, were measured in terms of local

standards at the field site which were provided and calibrated by

the Electrical Standards and Development Laboratory, U.S. Army

TMDE Support Group. All NBS transport standards were calibrated

by NBS before and after each trip. At Redstone measurements were

performed using the procedures described in this paper. The

results of all measurements described in this paper are expressed

as the differences, d.d, in parts per million, between the

measured values at RSA and NBS obtained from the equation

313
RSA ~ XNBS

x 10 = d.d ppm,

XNBS

where XRgA and XNBS are the measured values in terms of local

Army standards and the NBS standards respectively. In this paper

we will present an overview of the experiments, pertinent

results, and give some conclusions.

DC MEASUREMENTS

The NBS dc measurement package contained (1) a temperaturecontrolled standard cell enclosure with four cells, (2) a voltage

divider, (3) standard resistors, (4) six-digit digital

multimeters (DMM), and (5) auxilary equipment which included a desk-top computer, low thermal leads, and switches. The DMMs had

0.1 microvolt resolution, an input impedance in excess of 100

megohms, were IEEE-488 bus compatible, and were used as the

measuring instrument for all dc measurements at Redstone. In

order to minimize the effect of parasitic emfs in the measuring

instruments and some of the leads two observations were taken,

one with the DMM connected normally and the other with its input

leads reversed. Typically the observed differences rarely

exceeded 2 ppm. At NBS the linearity of the DMMs was determined

to be approximately within ±3 ppm over the full scale. For the

lower three digits the linearity was within ±1 ppm or less.

DC VOLTAGE : Two different experiments were conducted to

determine dc voltage. The first was a direct intercompar ison of

the RSA local voltage standard and the NBS transport standard,

the other a comparison of two measuring systems at nominally 1,

10, and 100 volts. Using standard Volt Transfer Program

techniques (VTP) the four cells in the RSA standard cell

enclosure were directly compared to the NBS transport standard

using a DMM to measure the differences between the cells

connected in series opposition. From the measurements at

Redstone and NBS the following results were obtained:

March 1982 ^RSA " ^NBS = °« 8 PPm

May 1982 E RgA - enbs =0.7 ppm.

The observed differences are consistent with those of the normal

VTP even though the settling time was short and only one set of

measurements were made at Redstone. The estimated uncertainty at

the three standard deviation level of this experiment is no

greater than 1.5 ppm.

The second experiment consisted of RSA and NBS making concurrent

voltage measurements using stable voltage sources on several

days. RSA and NBS each used a precision voltage divider and a DMM as the measuring system. The voltage dividers were connected

to the source and the divider output (at the 1 volt level) was

314
measured with a DMM having 1 ppm resolution. Using their

respective standard cells as the reference the RSA and NBS DMMs

were calibrated in terms of the local units of voltage. RSA and

NBS each calculated a voltage in terms of their respective

standards and the differences in these calculated voltages are

summarized below.

Nominal Difference (ERSA - ENBS ) in ppm

Voltage

(V) Day 1 Day 2 Day 3 Mean

1 +0.2 +0.2 +0.7 +0.37

10 -3.1 -2.7 -0.7 -2.17

100 -0.5 -7.1 -3.1 -3.57

At the 1 volt level the results are consistent with the

differences from the cell comparison. At the other two voltages

there is clearly an offset which may be voltage dependent. This

technique is very simple, can be carried out quickly, and could

lend itself to automation.

DC CURRENT ; A traditional approach to measuring dc current was

used and consisted of simply measuring the voltage drop across

calibrated resistors mounted in a temperature controlled oil

bath. RSA and NBS each concurrently measured nominal currents at

the 10 A, 0.1 A and 0.0001 A levels in terms of their local

standards (resistance and voltage). Current measurements were

made in March and May using several stable current sources. The

results of these measurements are summarized below.

10 A level I RSA - I NBS = +42 ppm

0.1 A level I RSA ~ I NBS = +9 ppm

0.0001 A level J RSA ~ I NBS

= +14 ppm

The large difference at the the 10 A is attributable to the

self-heating of the 0.1 ohm resistors. The difference at 0.1 A,

in as far as can be determined at this time, was mostly caused by

thermals at the potential terminals and temperature gradients in

the resistor caused by problems with temperature control and oil

level. At the 0.0001 A level a 10 k ohm resistor was used as the

shunt. From other work we concluded that leakage problems in the

oil bath could cause differences of the magnitude observed.

Certainly if more accurate current measurements are required then

self heating, leakage, and thermal problems must be addressed.

315
DC RESISTANCE : Resistance intercompar isons were conducted at the

1 ohm, 10 k ohm, and 10 M ohm levels. The NBS transport

standards were two Rosa type 1 ohm, two Rosa type 10 k ohm, and

two NBS constructed 10 M ohm resistors which were intercompared

with the RSA local standards. All resistors except the two 10 M

ohm were in an oil bath operated at a nominal temperature of 25

deg. C. At the one ohm level the resistors were measured by

applying a constant current of 100 mA to the three resistors

connected in series (one RSA and two NBS) and measuring the

voltage drop with the DMM. For the other two values the DMM was

used in the ohmmeter mode. In each case the value of the

transport standards was calculated from the relationship

R

x

= WE

s> where the subscripts s and x designate the RSA standards and the

NBS transport standard, respectively, E the DMM reading, and R

the resistance. The May measurements were performed when the

bath temperature sometimes exceeded 25 deg. C. The results are

summarized below.

Nominal RRSA _ in ppm

Resistance March 1982 May 1982

1 ohm +0. 2 +4.

10 K ohm -0. 6 -0.

10 M ohm +39 +58

During March, measurements were made when temperature conditions

were relatively stable there was little variability among the

runs, and the measurement process was well behaved. In May,

temperature and humidity conditions were far more variable and

there was much more variability in the results. In all cases the

measured differences were consistent with the estimated

uncertainties for the measurements made. The difference between

the two NBS transport standards, at the 10 M ohm level, when

measured using the DMM, at both Redstone and NBS, consistently

differed from those obtained by the normal NBS calibration

process.

AC MEASUREMENTS

AC VOLTAGE : The NBS transport standard was an unmodified

commerical ac-dc thermal transfer standard which was hand-carried

between NBS and Redstone. Two types of measurements were made at

Redstone: (1) the direct measurement of the ac-dc differences of

the transport standard and (2) a comparison of the Army field

site measuring system and the NBS transport standard. In both

March and May, the ac-dc differences of the transfer standard

were measured directly in terms of the Army standards in their

316
standards laboratory rather than at the field site. Because of

time restrictions it was necessary to limit the number of

measurements of ac-dc differences to two voltage levels, 5 and

500 volts, at each of three frequencies, 60, 400 and 6000 Hz. At

NBS , the transfer standard was also measured at the same voltages

and frequencies before and after transport. The results are

summarized below in the form:

DRSA " DNBS

where the D's are the mean ac-dc differences measured by the

respective laboratories.

Transport Date

DRSA " DNBS std - Dev * Standard ppm ppm

NBS March 1982 + 2.7 6.3

RSA* March 1982 -5.0 2.2

NBS May 1982 +2.0 5.6

* In March the Army also measured their own transport

standard and it was hand carried to NBS where it was

also measured.

The mean of the three is -0.1 ppm and the pooled standard

deviation is 5.0 ppm. The agreement of ac-dc difference is

remarkable considering the fact that the NBS uncertainty

assignment for this type of calibration is 50 ppm. Furthermore

the NBS transport standard remained quite stable over the whole

period of the work. These results suggest that a MAP service for

ac-dc difference might easily be developed.

The May comparisons at the field site of the RSA measuring system

and the NBS transport standard were performed at the same

voltages and frequencies as were used in March. However, the May

observations included measurements at the 50 volt level.

Measurements were made as nearly concurrently as practical using

the two systems and stable ac and dc voltage sources. After

applying the appropriate corrections to the measurements the

difference

RSA VNBS

was calculated for each measured value. The mean difference for

all voltage observations was +23.2 ppm and the standard deviation

of a single observation was 7.5 ppm. Although well within the

NBS assigned uncertainty of 50 ppm the observed difference is

larger than can be accounted for by the standard deviation of the

measurements made. It was not possible, within the time

available, to investigate the the orgin of this difference.

317
AC CURRENT: The measurement of ac current was performed using

the ac-dc transfer standard and three current shunts designed to

operate with the unit. In May the transport standard and the

3 A, 0.1 A, and .01 A shunts were calibrated as a unit in terms

of the Army primary standards . Except for the 3 A shunt the

applied current was the rated shunt current. In the 3 A case,

the Army used 2 A and NBS used 3 A. The mean difference between

the Army and NBS corrections was +7.7 ppm and the standard

deviation was 8.2 ppm. Although the uncertainty is somewhat

larger than than obtained for voltage, these results are in good

agreement with the voltage data.

Comparisons of the measurement system at the field site and the

NBS transport standard were conducted in both March and May. The

mean difference in the measured currents was -3 ppm with a range

of 58 ppm. Again time constraints prevented further

investigation of the sources of the observed differences.

However, it was observed that certain of the equipment was very

temperature sensitive. Since it was designed to operate in a standards laboratory environment, this observation is not

surprising.

AUTOMATED MEASUREMENTS AND DATA PROCESSING

Automation of the NBS electrical measurements described in the

preceeding sections provided a logical and practical means of

increasing the efficiency of taking measurement data.

Specifically, automation afforded several advantages over manual

methods. First, automation improved the integrity of the data

taken. Since the data was in machine-compatible form, it was

possible to record, print and analyze the measurement results

during the measurement process. This eliminated the manual data

recording process in which errors are highly likely. Second,

automation increased the speed at which data was taken. The NBS

experience in two field tests at RSA consisted of performing

similar tests both with and without the aid of an automated

system. The automated measurements permitted much more data to

be taken in approximately 30 percent less time. This increased

efficiency and allowed more measurements to be made, thus

increasing the experimenter's confidence in their data. The

third benefit that was realized by automation was the consistency

of performance of a measurement. Each time the measurement

program was run, the same action took place. In our case, the

measurement procedure coded into the software was simple. It

consisted of the commands to set a digital mulitmeter to the

desired function, to wait for a specified length of time, and to

print the results. By having these commands in software, errors

introduced by the operator in setting up and using the instrument

and in the recording the data were virtually eliminated. The

final advantage of automation realized in this work was to

provide a documentation of the procedures used to make the

measurements. The software provided an unambiguous record of the

sequence of events that occured during the

318
measurement process. This may be valuable, especially in field

tests such as reported here here, where further data analysis

depends on the knowledge of the details of measurements made.

In fairness, it should be pointed out that there were some

disadvantages to automating the measurements. Foremost, care

must be exercised to assure that the software operates properly.

This was accomplished by writing software in modular form, proper

documentation throughout the software with remarks and comments

on what was being calculated, and proper testing of the software

under all conditions of data input.

For the implementation of automation techniques in measurement

systems, it is desirable that the test instrumentation be easily

interfaced to the controller being used. The approach used in

this work was to use an IEEE-488 (1978) interface among all

instruments. The IEEE-488 bus is a "party-line" bus in which

instruments are attached in parallel to all the wires of the bus

all of the time. Thus, only one instrument can be sending data

at any given moment. Software in the controller determines the

sequence of messages that will be transferred on the bus. In the

NBS experiments, the most used software program was a simple

routine, written in BASIC, which issued instructions to the

digital mulitmeter to make ten readings average the readings, and

print the average, standard deviation of the measurements, and

the highest and lowest readings in the group of ten.

Additionally, a time was printed which could establish the order

of of the measurements should the need arise. The results were

printed on a thermal-paper printer integral to the controller.

These records then provided the raw data that was further

evaluated using the controller as a simple calculator. By

processing the data promptly, a potentially serious problem with

one of the NBS procedures was discovered immediately. This

procedure was then modified, the measurements repeated, and

satisfactory results were obtained. The next logical step to

further improve the system described would be to record the

measurement data directly onto magnetic tape or disk files and to

immediately process the data and display the corrected results.

Such a procedure would give virtually real-time information on

the calibration of complex equipment.

SUMMARY AND CONCLUSIONS

For the most part no uncertainties have been assigned to the

particular measurements because of the complexity of the overall

measurement processes involved. Any assigned uncertainty must

include an error analysis and assignment of uncertainties to the

NBS transport standards, the NBS measurements of the Army primary

standards, the Army surveillance process, the local Army

standards at the field site, variability of the voltage and

current source, effects of the local environment on the

319
measurement process, and, above all, the measurement processes

used by both the Army and NBS for this work. As of this writing

the differences reported seem to be consistent with the

preliminary estimates of the uncertainties for the various

measurements. It is heartening to note that we were able to

conduct a series of measurements between two metrology

laboratories and obtain the quality of agreement reported in this

paper. In closing, the following tentative conclusions have been

drawn based on this work.

There is a need to develop standards that can operate in the type

of environment that was encountered in this work.

In the area of ac measurements there is a need to develop tranfer

standards that will reduce the time required to make the

necessary measurements. It was found that the ac measurements

were the most labor intensive of all the measurements made.

It is essential that calibrations of the type described in this

paper be carried out in real time so that problems can be

resolved as they occur rather than simply be identified after the

fact. NBS experience in these experiments indicates that the

microcomputers are useful for this purpose because they provide

on-site data analysis and may be used for control of the

calibration process.

Finally, this work, even though limited, shows that (a) field

calibrations are one alternative to returning the equipment to

the standards laboratory; and (b) that such calibrations can be

carried out to a high accuracy and over a broad range of values

for different electrical quantities using a relatively small

number of standards and instruments.

320
68-608

TRANSFER OF THE UNIT OF VOLTAGE

Presented at the 1968 ISA Annual Conference and Exhibit

October 28-31, 1968

NEW YORK CITY

Copyright 1968

INSTRUMENT SOCIETY of AMERICA

530 William Penn Place

Pittsburgh, Pennsylvania 15219

321
68-608

TRANSFER OF THE UNIT OF VOLTAGE

Norman B. Belecki

Physicist

2802 Inertial Guidance

Newark, Ohio

ABSTRACT

In January 19°7 a joint project was established

between the United States Air Force Calibration

and Metrology Division and the National Bureau

of Standards to investigate the problem of transferring the unit of voltage at the highest

accuracy levels. After ascertaining the relative

stability of the voltage standards in the two

laboratories various types of transfer devices were evaluated including saturated and unsaturated

standard cells and Zener diodes in various environments . Statistical measurement schemes were

employed to eliminate average residual emf 's in

the measurement system and to provide estimates

of the system precision. The details and results

of this project will be discussed at length.

INTRODUCTION

Starting in the first month of 1967, the USAF

Calibration and Metrology Division and the

National Bureau of Standards undertook a joint

project to study the problems associated with

the inter-laboratory transfer of the unit of

voltage. The primary objectives of the study

were these:

(a) To reduce the uncertainty of assignment

of the unit of voltage

(b) To reduce the time, and hence the expense, required to make an assignment

(c) To develop and use statistical procedures

for maintaining surveillance over the unit In order to pursue these objectives, every aspect

of the problem had to be considered. Various

types of transportation were considered. Techniques were developed for the handling and pro- cessing of the data taken.

STATISTICAL DESIGNS

Statistical measurement designs were used throughout the program. The solutions of such designs

give one the value of each cell relative to the mean value of any cell or cells within the group,

the standard deviation of a single measurement,

the mean left-right component, and the difference

of each observation from the value predicted by

the values of the cells. Physically, the leftright component can be a detectable thermal emf

within the measuring circuit or a scale shift in

the potentiometer used to measure the difference

and Calibration Group

between two cells . Components of the standard

deviation of an observation are:

(a) Short term instability of the cell emf 's (b) Lack of perfect linearity of the measuring

device

(c) Lack of infinite resolution of the measuring device

(d) Instability of the left-right component

with time

These designs were suggested for use by Mr. Joseph

Cameron, Chief of the Statistical Engineering

Section of N.B.S., and were used throughout the program. One is for the comparison of a group of

nine cells with a group of three in three sets of

eighteen observations each. The other two designs

are used to intercompare a group of six cells with

a group of four or another group of six. Each of

these designs requires twenty four observations.

Data garnered in these designs is processed by the

N.B.S. on a time shared computer using a program

written by Mr. Eicke of the Electro-chemistry

Section of N.B.S.

DEFINITION OF LOCAL STANDARDS

For the purposes of this experiment, local stan- dards were defined both at N.B.S. and NAFS. At

the N.B.S., the nine N.B.S. made cells of the

Secondary National Reference Group defined as the

local standard are housed in a center N.B.S. temperature mineral oil bath maintained at 20°C.

This bath has a thermistatic (on-off) controller

and has a short term temperature stability of about .01°C. Differences in emf of the various

cells are measured using a commercial standard cell

comparator and electronic null detector. The reso- lution is 0.1 microvolts.

In the USAF Primary Electrical Standards at Newark

Air Force Station, Ohio, the local standard was defined to be nine commercial saturated standard

cells . These cells are contained in a commercial

oil bath maintained at 35°C The temperature con- trol unit has proportional, rate, and reset capa- bilities. The short term temperature stability is of the order of 0.0005°C. Due to problems created

by the large difference between the bath temperature and ambient (25°+ .5°) temperature, the cells

are permanently connected to a mercury cup switch

located in an oil filled aluminum box. This box

is contained in a larger aluminum box located in

322
an instrument

console beside the oil bath. The boxes, leads,

and the conduit through which the leads run form

the basis of a guarded, shielded measuring system.

Leads are provided in this system for seventeen

three cell enclosures and one four-cell enclosure.

Enclosures are attached to the system upon arrival

and remain physically undisturbed throughout the

tests. Differences between cell emf 's are measured using a potentiometer of 0.1 microvolt reso- lution. Null Detection is by means of a photocell

amplifier which drives a mechanical galvanometer.

STANDARD CELL TRANSPORT UNITS

A representative sample of commercially available

portable standard cell enclosuies were utilized in

this program. Preliminary studies were also made

of Zener diode units and unsaturated cells

.

carrying by an aircraft, hand carrying by auto- mobile (using the car's electrical system), and by

air freight. All Three of these methods were

utilized. For the third, arrangements were made

with one of the major airlines for a monitored

shipment by reserve air freight in a heated cargo

space. Special fiberglas packing cases with polyurethene foam solution around the enclosure and its

batteries were procured. The entire transportation

process was tightly scheduled with the enclosure

held at the airport of the destination for personal

pickup by laboratory personnel. In transfers be- tween NAFS and the N.B.S., this system worked well

at all times despite inadequate instructions

given the carrier on one occasion. In cases involving a change of aircraft enroute, some difficulties were encountered. Evidently, the

enclosures received a severe beating during the

transfer.

The table shows the types of enclosures used.

Temperature fluctuations could be monitored in

each enclosure by means of a thermister bridge

with a readout dial on the face of the device. In

addition, one of the thirty degree baths and both

of the twenty eight degree baths have provisions

made for the insertion of a platinum resistance

thermometer. All units have provisions for a battery standby power supply. The twelve cell

28 degree bath was a modification of a commercial

bath. There was a massive aluminum block added

for thermal bulk, and four groups of three cells

each were installed with a guard circuit. There were two groups of portable saturated cells and two groups of shippable unsaturated cells . The

modification was unowrkable due to the lack of

geometric symmetry of the block and the location

of the control sensor. The four cell 35 degree

bath was modified because of leakage to the oven from the cell leads. The modification consisted

of TFE sleeved, pure copper leads with a driven

guard.

TRANSFERRING THE UNIT

It was decided at the beginning of the program

that the best method to handle the information was

to monitor the differences in assignments to transport units of the two laboratories. The resultant

plot (voltage differences vs. time) would be a band ideally centered about zero with a zero shape.

The distribution of the points about the center

curve of the band and the shape of the band itself

determine the final accuracy of assignment. As long as the points added to this plot remain within the acceptable bounds from the center curve, the same value is used for the mean of the local

cells. The process is then said to be in a state

of statistical control. If, however, a point is

plotted which is out of bounds, the process is then

said to be out of control. At this point, some sort of validation of the out of control point is

required. Unless there is obvious reason to believe the transport unit is faulty, the transfer TABLE I. Types of Transport Units

No. of No. of No. of Norn. Cell Type of Typical

Enclosures Cell a Ovens Temp. Temp . Cont

.

Short Term

Stab . *

2 3 2 35 Proportional .4 - .5

2 k 2 30 Proportional .2 - .3

1 3 2 32 Proportional .3 - .h

1 12 (6 Unsat.) 1 28 Proportional .j - 1.0

1 8 (4 Unsat.) 1 28 On-Off .5 - .8

1 in 2 35 Proportional .3 - .k

* Average per cell 3 sigma for two weeks readings

In view of the second objective of the program, it was necessary to maintain the temperature integrity of the enclosures at all times. This pre- cludes all modes of transportation but three, due to the bulk of batteries. The three are hand

must be accomplished again. If subsequent transfers are also out of control, of course, the former

statement of uncertainty no longer applies, and

either a new local value is required or a new accuracy statement, or both.

323
TABLE II. NAFS Standard Disturbances

Date Occurrence

25 May 1967 Mechanical failure, Loss of temperature

22 Jun 1967 Power Failure, 5 Deg. loss in temperature.

9 Aug. 19o7 Electrical difficulty in measuring circuit, at mercury cups.

7 Oct. 1967 Power failure, Loss of temperature; Cells placed in

aluminum cans in bath.

17 Nov 1967 Cells lowered in oil bath.

22 Dec 1967 Mylar sleeves placed in aluminum cans

22 Jan 1968 Emergency power system installed, 0.1 Deg. loss

temperature

12 Feb 1968 Removed excess cans from bath; Sleeved positive

leads with TFE.

2d Apr 19&8 Excess items removed from bath.

The value of the local standard at N.B.S. was determined by inter-comparing the nine cells with

cells from the National Reference Group. Figure

One shows the value of the mean of the group during the first fourteen months of the experiment.

The calibration points are shown as data. However,

it is important to note that the average mean value

as represented by the solid line was in all cases taken to be the value of the group mean. As seen

in the seventh month, the cell responsible for the

downward shift of the mean emf was removed and

another substituted. From this point, the assign- ment has remained constant.

The average value of the group of nine in the Air

Force laboratory was assigned as the result of two interchanges during which two portable standard

cell enclosures were carefully hand carried be- tween Gaithersburg and Newark while under power

and or prior data taken on these two enclosures

(the first enclosures in Table One).

At this point, the use of the various types of

transport devices was begun. The graph (Figure

Two) shows the control chart kept for the first

sixteen months of the program. For the first

fourteen months of the project, the points

(representing complete transfers) are within con- trol. One of the observations made is that

Box 100 (represented by crosses) always seems to

have a lower difference in assignment than

Box 200 (represented by the dots). A similar

relationship exists between Boxes 1100 and 1200

(the boxes represented by squares and triangles).

After the fourteenth month, the entire picture is

shown out of control as the result of some changes

made in the bath containing 'the nine cells.

The local standard at Newark was disturbed a number of times during the experiment. Table Two

lists the major difficulties encountered in the

system during the experiment. As the October

power failure resulted in the return of the cells to room temperature, it was decided to mount them

in aluminum cans to remove the effects of oil

turbulence. The resulting recovery to stability

of the mean of these cells is shown in Figure

Three. It is interesting to note that internal

differences alone would have indicated stability

in two to three weeks during which time the act- ual mean value would have been high in excess of

a microvolt. On the other hand, the internal

differences indicated some types of disturbance

at about sixty-five days. At that time, the mean

was relatively stable. This indicates the 4'olly

of relying on internal measurements alone to determine the validity of a standard, especially if

all of the components are similar and in the same environment.

THE I.B.M. - NAFS - N.B.S. TRANSFER

Two additional experiments were attempted involving

a third laboratory, that of I.B.M. in San Jose,

California. The two thirty degree enclosures were used, one being shipped to NAFS and the

other to I.E.M. from N.B.S. At the end of a two week period of testing, the laboratories exchanged

boxes. The end of another two week test period

saw both boxes returned to N.B.S. As shown in

Table Three, the enclosures each experienced a real change in emf during the interchange between

I.B.M. and NAFS.

It is to be noted that all shipments to I.B.M. and the return trip from I.B.M. to Gaithersburg

involved power failures of the two enclosures.

It was also noticed that the enclosures received

at NAFS were at a higher temperature than nominal,

undoubtedly due to the extra insulation afforded

324
Mean of Mine N.B.S. Cells

0 1 2 3

'** 5 6 7 3 9

T^tae, Months

Figure One

"lO H S 13* W

1.0

0.8

0.6

0.4

0.2

0

-0.2

-0.4

-0.6

-0.8

-1.0

NAFS Control Chart

Upper Control Limit

o o

Lover Control Limit

12 3 4 5 6 7 8 9 10 11 12 13 14 15 16 Tine (Months)

Figure Two

THIRTY FIVE rEGRSE

BATH RECOVERY

-1

40 60 80 120 litO

TIKE (Days)

FIGURE THREE

325
TABLE III. Interlaboratory Differences

Enclosure Difference in Microvolts

1100 V(NBS 1) - V(NBS 2) = -1.2

V(NBS 1) - V(IBM) = +.3

V(NBS 2)-- V(NAFS) = -.7

1200 V(NBS 1) - V(NBS) = +1.8

V(NBS 1) - v(nafs) = -.3

V(NBS 2) - v(ibm) = -.9

by the packing case.

The experiment was repeated at a later date. The

results of the second experiment were inconclusive

due to extreme mishandling of the enclosures again

between I.B.M. and Newark , and due to problems

in the measurement system at Newark. In this

instance, physical damage was actually done to the

shipping cases of one enclosure. This experiment

will be repeated, soliciting extra care on the

part of the carrier and involving at least one enclosure at thirty five degrees.

UNSATURATED CKT.T.S AS TRANSFER DEVICES

The use of unsaturated cells for the transfer of

the unit was partially investigated. Each of the two twenty eight degree baths contain unsatu- rated cells. Although the twelve cell bath was a failure, preliminary tests on the eight cell

enclosure indicate that the standard deviation of the measured points from the predicted emf at any time was 0.3 microvolts, indicating that these

cells may be potentially used to transfer the unit

with an uncertainty of one microvolt. This information assumes a least square fit of a straight

line through the data taken during the duration

of a normal transfer. A further test of the use

of unsaturated cells is currently under way be- tween the two laboratories and other tests are planned by the USAF.

ZENER DIODES

maintaining, and proving the uncertainty left to

the using laboratory.

Thirty five degrees centigrade is too high a temperature for the maintenance of cells to be used

on a permanent local standard. The recovery time

from temperature loss is sixty five days as com- pared to about a week or two for cells at 28°C.

The standard deviation for a single observation

was shown to be 0.08 microvolts at Newark Air

Force Station while it is 0.18 microvolts at the

N.B.S. The increased precision at NAFS is sur- mised due to the superior temperature control made

possible by the proportional, rate, and reset

controller.

Standard cell enclosures which operate at higher

temperatures are seen to be effected less by the

variety of ambient conditions encountered in a

transfer. This was illustrated by the overheating

of the two thirty degree boxes in transit.

The recovery time for saturated cell enclosures

transported maintaining temperature integrity is of

the order of a few hours. Readings taken immediately upon the arrival of the enclosure generally

tend to be four or five tenths of a microvolt

from their final value. Data taken twenty four

hours after arrival usually is within 0.1 microvolts of the average value.

ACKNOWLEDGEMENTS

Preliminary tests have also been made on the use

of self contained Zener diode packages . It has been shown that the emf of the units at any time

is predictable to within about three microvolts

without a transfer. One transfer has been made

utilizing eight Zener diodes . The results have

been varied enough to permit no firm conclusions. More work will be undertaken in the future.

RESULTS AND CONCLUSIONS

The author wishes to acknowledge Henry Ellis and

Ann O'Toole of N.B.S. and John Donahue and Gerald

Howard of NAFS for making the observations upon

which this paper is based. Thanks of the author

go to the I.B.M. standards laboratory and its

chief, Mr. DeWayne Sharpe, for their participation.

Without the efforts of Mr. Woodward G. Eicke of N.B.S. in handling the logistics, data, and many

of the details, this program would not have been

the success it was.

The primary unit of D.C. voltage of the USAF has

been certified to have an uncertainty of 0.5

microvolts at Newark Air Force Station. Formerly,

cells were certified accurate to one microvolt at N.B.S. with the responsibility for transporting,

KEY WORDS

Saturated Standard Cell, Unsaturated Standard Cell, Zener Diode, Standard Cell Enclosure, Standard

Deviation, Control Charts

USCOMM-NBS-OC 326
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-23, NO. 4, DECEMBER 1974

Regional Maintenance of the Volt Using NBS

Volt Transfer Techniques

WOODWARD G. EICKE, JR., member, ieee, and LAUREL M. AUXIER

Abstract—In cooperation with five industrial standards laboratories, the National Bureau of Standards (NBS) studied the feasibility

of establishing a regional volt maintenance and surveillance program

in the greater Los Angeles, Calif., area. The objectives were to improve surveillance, to reduce dependence on NBS, and to have each

laboratory maintain its unit of voltage to within 1 ppm of the U.S.

legal volt. A two-phase program was established, the first to characterize the five laboratories and the second to carry out the surveillance. After 3 years all laboratories were found to be maintaining

their unit to within better than 1 ppm of the U.S. legal volt.

I. INTRODUCTION

THE U.S. legal volt is disseminated by the National Bureau of Standards (NBS) to several hundred pri- vate and public facilities throughout the United States via

either unsaturated or saturated standard cells submitted

by NBS clients for calibration. In late 1967 it became

obvious that the services offered did not meet the accuracy

requirements of a small but important group of users. To

alleviate this problem, NBS, in cooperation with the U.S. Air Force, undertook a program to improve the inter- laboratory transfer of the unit of voltage. As a result of

this work a new service, the NBS Volt Transfer Program

(VTP), was developed and made fully operational in early 1971 [1], [_2~\. This technique reduced the uncertainty of the transfer of the unit of voltage from approximately 2 ppm using the normal methods to 0.4 ppm.

In early 1970 the authors discussed the possibility of extending the VTP technique to the establishment of a

regional volt surveillance program in the Los Angeles,

Calif., area with the goals of improving local volt sur- veillance and reducing the dependence of the participating laboratories on NBS. As a result, five industrial laboratories in the greater Los Angeles area established the Southern California Standard Cell Interchange Program

(SCSCIP) to work with NBS on a study of the feasibility of setting up a regional volt maintenance program. Participating were the Autonetics Division of Rockwell

International, Inc., Beckman Instruments, Inc., Electrical Standards Repair Service, Inc., Lockeed-California Company, and TRW Systems Group. Since no one had pre- viously undertaken such an endeavor, a two-phase proManuscript received July 3, 1974; revised August 21, 1974. This paper is a contribution of the National Bureau of Standards and

not subject to copyright. W. G. Eicke, Jr., is with the Absolute Measurements Section,

Electricity Division, National Bureau of Standards, Washington,

D. C. 20234.

L. M. Auxier is with Beckman Instruments, Inc., Fullerton,

Calif.

gram was formulated first to establish the feasibility of such a program, and second to develop an operational

plan to carry out, on a continuing basis, a regional volt maintenance program. After discussions with the participating laboratories we set, as a goal, 1 part in 106 as the maximum difference between any participant's as-maintained unit and the U.S. legal volt.

II. FEASIBILITY STUDY

.4. Experiment Plan

Each of the five laboratories maintained their local volt via saturated standard cells, kept under close temperature control, and each intercompared its cells using

the series opposition method. However, no two laboratories were identical; some maintained their reference cells in

oil baths, while others used temperature-controlled "air baths." In addition, instrumentation, operating procedures, and stastistical techniques also varied among the

participants. It was clear from a study of the operating

procedures of the various laboratories that all were comparable; hence, no major changes in their method for maintaining their volt were required at that time. Therefore, it was decided that as-maintained units of voltage would be best compared using the A

T BS VTP techniques. Furthermore, since each laboratory had a trans- port standard available, five transport standards would be

employed simultaneously. In this way the desired meas- urements of the differences between as-maintained units

of voltage could be carried out in 12-15 weeks. The design

of the experiment showing the five laboratories, designated

A through E, and the ten transfers is depicted in Fig. 1. The arrows indicate the laboratory supplying the transport

standard for each path. For example, for path 1, laboratory

A sends its transport standard to laboratory B. All the data were sent to NBS for processing and analysis as they were obtained.

The actual technique used to measure the differences in the units is essentially that reported in [1] and [2 J. Briefly, the home laboratory calibrates its transport

standard in terms of its as-maintained unit of voltage VH and then transports the enclosure under power to the

designated laboratory. There, it is calibrated in terms of that laboratory's as-maintained unit Flab- Finally, it is returned to the home laboratory where it is recalibrated.

If Ey/d) and Ehm are the assigned EMF's at the home

laboratory in terms of VH, and .Blab is the EMF in terms of Vlab at the receiving laboratory, then the difference in

327
K1CKK AND AUXIKR: NBS VOLT TRANSFKR TKCHNIQU F.S

Fig. 1. Design for Phase I experiment. Arrows indicate the labora- tory furnishing the transport standard for that leg of the experi- ment.

REFERENCE STANDARDS

P A

0 R

R D

1 2 3 N ! N + + +

+ +

- + + +

1 + +

T S

Fig. 2. Experiment design used to intereompare reference stand- ards and transport standard. Signs indicate relative position in the measuring circuit.

the as-maintained units is

V V - I^'Ilab - I E\„

V LAB — V n = — I ii ( 1 ) \E} H where the \E\'s are the numerical values only of the EAIF's assigned by a given laboratory and

EH = i(Eam + EH(2) ). (2) The timing of the experiment was such that any linear

drift in the EMF of the cells in the transport standard was

automatically eliminated from the final result given by ( 1)

The procedures used to intereompare the transport

standards and the reference standards were those used in the VTP, and the experiment designs were those shown in Fig. 2, where the signs + and — indicate the position in the circuit. All transport standards contained either three

or four cells, and intercomparisons were made between the

reference and transport standards 2 to 4 times per week,

with measurements extending over a 2- to 4-week period.

B. Experimental Results The experiment was begun on November 9, 1970, and

completed February 15, 1971, according to the schedule of Table I. The planned schedule called for completing the experiment by January 22, 1971, but several minor problems including the Los Angeles earthquake caused slippages

of a few days each, while the Christmas holiday caused a 1-week delay It was also found that approximately 3 weeks at each stage yielded much better results, because a number of enclosures showed some instability for the first week, even though they were carefully transported under

power less than 80 km by automobile. Results from the

individual daily runs were combined and the average EMF

of the cells in the transport standard for each stage was

calculated. These results are summarized in Table II. Although the data of column 10, Table II, are in terms

of the home laboratory Vu. we can view the data as being

referenced to a single unit of voltage because the inter- laboratory differences are small. This single unit is taken as the unit of voltage for SCSCIP, Vsc . We, therefore, define Vsc as the mean unit of voltage for the five laboratories. Now the difference between the ith laboratory's unit V\

and VSc can be readily calculated using the method of

least squares. Using the procedure for 5 cells given by

Eicke and Cameron [4] one obtains the results presented

in Table III. From the data of Table III the predicted observations

and deviations (observed minus predicted) were calculated (columns 11 and 12 of Table II). In addition, the lack of closure was estimated to be —0.09 ^V as shown at the bot- tom of column 10. The standard deviation of a single differ- ence was found to be 0.35 /iV for 5 degrees of freedom. Laboratory A had also participated in two NBS VTP

experiments that bracketed this work. The data showed

their unit of voltage was constant to within the experimental

error of the VTP and had been adjusted to make VA —

Fnbs = 0 to within experimental error. Therefore, Fsc was

set equal to Fnbs and the results are shown in the second

column of Table III. From the individual designs carried out by each laboratory, one obtains the left-right component, an estimate of the precision of a single observation, the precision of the assignment of the transport standard

in terms of the reference group, and some information on

individual transport standards. This information is also summarized in the last three columns of Table III. The

left-right component is an estimate of the residual constant EMF's in the measuring circuit. In two cases they were on the order of 0.1 to 0.2 /uV. Since they remained

constant during any run their effects on the calculated data are eliminated by means of the experiment design. For

each laboratory, the standard deviation of a single obser- vation is given in the next to the last column. Each figure

is an estimate of the precision of the laboratories measure- ment process and most are comparable to data obtained

from the regular VTP [4]. The last column gives infor- mation on the day-to-day performance of the process. Based on these data, participants were able to make a number of minor changes in their process that improved

their overall capability. Using the data in the third column

of Table III, the units were all adjusted to make Flab —

Fnus = 0 to within experimental error.

III. PHASE II. ROUTINE REGIONAL

SURVEILLANCE

Because the Phase I program was far too complex and

time consuming to be used for routine surveillance, a

simpler method had to be devised. The obvious method for setting up a routine program was to use one laboratory as a pivot as shown in Fig. 3. Using this scheme, each labora- tory submits a calibrated transport standard to the pivot

328
IKEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1974 TABLE I Schedule for Phase I Experiment

Transport Standard

Calculated at

Designated Numer of From To Home Laboratory Days

November 9, 1970 November 20, 1970 x 12 November 21, 1970 December 11, 1970 X 20 December 12, 1970 January 8, 1971 X 27 January 9, 1971 January 29, 1971 X 21 January 30, 1971 February 15, 1971 X 16

Note: Average time at each laboratory was about 3 weeks.

TABLE II Summary of Phase I Interchange

EMF's

Path Reduced by Labora- Number (jiV) tory Emu

G»V»)

Home Laboratory

(mV»)

Assumed Mean*

(mV„)

Standard Deviation of Assumed Mean

Receiving

Laboratory

Laboratory ID

Slab

(mV lab )

Difference in Units

Predicted Flab — Vh Deviation (mV„) (mV) (m V)

1 1018100 A 37. 35 37. 38 37 36

2 1017600 B 93. 34 95 99 94 66 b 3 1018100 C 36. 13 36 05 36 09

4 1017600 D 93, 01 93 11 93 06

5 1018100 E 44 75 45 31 45 03

6 1018100 A 37 38 37 46 37 42

7 1017600 B 95 99 97 51 96 75 b 8 1018100 C 36 05 36 06 36 05

9 1018100 D 93 11 93 45 93 28

10 1017600 E 45 31 45 .13 45 .22

0. 015 B 36. 03 + 1.31 + 1 .57 -0.26 C 96. 56 -1.86 -1.22 -0.64

0. 040 D 35 65 +0.43 +0.32 +0.11

0. 050 E 93 58 -0.51 -0.77 +0.26

(i 280 A 45 28 -0.24 -0.17 -0.07

0. 040 C 36 86 +0.55 + 0.26 +0.29 D 97 49 -0.73 -0.99 +0.26

0, 005 E 36 71 -0.64 -0.36 -0.28

0. 170 A 94 03 -0.74 -0.85 +0.11

0, 090 B 43 64 + 1 .55 + 1 .49 +0.06

8 = 0.123

Assumed mean at receiving laboratory.

b Cells in this enclosure showed a relatively linear drift. The assumed mean took this drift into account to eliminate confusion with Eho)-

c Standard deviation not calculated because of drift.

TABLE III Summary of Differences in As-Maintained Units and Other Pertinent Laboratory Data

Laboratory

Difference Flab — Fgc

(mVsc)

in Units Flab — Fnbs

(mVnbs)

Left-Right

Component

GuV)

Standard Deviation"

Single Observation

(mV)

Standard Deviation 1" Assignment Between Days

(mV) A -0.57 0 0.021 0.071 0.057 B + 1 .09 +1 .66 -0.017 0.131 0 . 082 C -0.22 +0.35 0.065 0.063 0.200 D +0.19 +0.76 0.161 0.09 0.098 E -0.49 +0.08 -0.028 0.23 0.183

a Standard deviation of a single observation corresponds to the within day standard deviation.

b This quantity is calculated from the individual runs.

laboratory; at the same time NBS supplies a transport

standard. The pivot laboratory intercompares the NBS

transport standard with its laboratory reference group and

all of the transport standards, a process requiring 3-4

weeks. All of the daily data are forwarded to NBS for processing. Upon completion of the measurements all transport standards are returned to their respective laboratories where they are again measured over a 2-4-week period. After all the data have been processed by NBS, results are analyzed and reports of the test are sent to each laboratory, which give the difference Flab-Fnbs, recommended

adjustments to the local unit, and other process data. The

pivot laboratory assignment is based only on the comparison between their reference group and the NBS trans- port standard. For each transfer a different pivot labora- tory was selected so that in the long run all laboratories

in SCSCIP are treated in the same manner. After each

experiment, if required, the units of voltage are adjusted to make Flab- Fnbs = 0 to within experimental error. To date a total of three such transfers have been carried out and the results are summarized along with other

pertinent data in Table IV. Lines 1 and 3 give results of regular transfers between laboratory A and NBS, and line 2 gives the normalized data from the Phase I experiment,

329
EICKE ANL) AUXILR: NBS VOLT TRANSFER TECHNIQUES

/

/

1

I

1

I

TEST REPORTS ^

NBS

PIVOT LAB - # *

K —i—j— j_ _t _^ _ _i

1 » I

LABS

Fig. 3. Phase II showing by means of large arrows the movement

of transport standards. Flow of data and results are also shown.

TABLE IV Summary of Phase II Experiments

Laboratory A Laboratory B

Pivot Ob- Ad- Ob- AdLaboratory Date served" justed 6 served" justed 1'

Laboratory C Laboratory D Laboratory E

Ob- Ad- Ob- Ad- Ob- Adserved" justed b served" justed b served" justed 6 SCSCIPC

A* July 1970 0 +0 80

A' Dec. 1970 -0 11) +0.52 -0.10 +0.52 AA Apr. 1971 -0 18 +0.31 C Oct. 1971 -0 3K -0.09 -0.55 -0.27 A Oct. 1972 -0 27 -0.27 +0.24 -0.31 D Nov. 1973 -0 05 -0.05 +0.22 -0.08

0.10 +0.52 -0.10 +0.52 -0.10 +0.52 -0.10

0.36 -0.08 -0.08 -0.20 +0.35 +0.63 -0.20

0.30 -0.66 -0.26 -0.16 +0.28 +0.07 -0.06

0.69 -1.35 -0.24 -0.40 0.00 -0.07 -0.15

" Observed difference experimentally determined.

b Difference in the units normalized to the Phase I experience.

c All units were adjusted to make Flab - Fnbs = 0 using the two VTP experiments with Laboratory A. It was assumed that Vji was

constant; however, later data showed a slight drift.

d Regular VTP transfers between NBS and Laboratory A.

while lines 4, .5, and 6 summarize the Phase II transfers from October 1970 to November 1973. The data of this table constitute a complete history of the SCSCIP data

going back to July 1970. For each laboratory two values

are given. The first is the actual measured difference be- tween the laboratory and NBS. The second is normalized

to December 1970 (except laboratory A whose base date is July 1970) , in order to show changes with respect

to time in the as-maintained units as a result of changes in the EMF's of the reference cells. The normalization takes

into account the drift in the U.S. legal volt prior to July

1972 and the adjustments to the local unit as the result of previous intercomparisons. Field et al. [o] reported that the U.S. legal volt as maintained using saturated cells drift ed at the rate of about —0.41 pY per year relative to the

assigned value of 2e/h from July 1971 to July 1972. Assuming this relationship was valid for the 12 months prior to July 1971, all data prior to July 1972 were adjusted accordingly. The last column of the table gives V*sc — FN bs- This

latter quantity is simply the mean of the 5 measured

differences. The observed differences represent, within

experimental error, the changes from one calibration to the next of the local units as preserved by means of saturated

cells. In some cases the actual constitution of the laboratories' reference group was changed between transfers.

For three of the five laboratories {A , C, and D) the normalized differences show definite time dependence, as is shown

in Fig. 4, which is typical of other VTP data [4]. In the

case of the other two laboratories no such dependence was

detected. Analysis of the data from individual designs and

changes in EMF of the transport standard as a result of transport indicated that the various processes were similar to those of the VTP. Therefore, the overall standard de- viation of a single transfer between each laboratory and

the pivot laboratory was taken as that of the NBS VTP,

or 0.14 nV. Since two transfers were involved the standard

deviation of a transfer between NBS and all but the pivot

laboratory is 0.20 nY.

IV. DISCUSSION OF RESULTS

The Phase I study clearly demonstrated the feasibility of establishing a relatively simple volt maintaince program

using the basic VTP techniques with only slight modification. Further, for the first time, to the authors' knowledge, it provided quantitative data on a closed loop ex- periment comparing several units of voltage. Phase II results showed a definite improvement in both the maintenance of the local units and the performance of the trans- port standards. It is our opinion that the repeat of the Phase I experiment would result in a standard deviation

330
IK EE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1974

+1.0

1970 1971 1972 1973 1974 TIME

Fig. 4. Behavior of the reference standards for the five laboratories showing the drift of their standards relative to the U. S. legal volt. (See text.)

substantially less than the observed 0.35 ixV. Even so, the

results are still in excellent agreement with those obtained

via the regular VTP. Furthermore, the Phase I technique can be readily extended to other size groups 'without un- duly increasing the amount of work to be done. The

Phase I experiment also identified problems heretofore unknown to the laboratory personnel, most being readily solved prior to starting Phase II. All of the laboratories had received their initial assign- ments from NBS using the conventional service which

required sending their standards to NBS for calibration.

It is heartening to note that the largest observed difference in units (1.66 /iV) was well within the expected range

(2 to 4 fiY) when allowances are made for long-term drift and transport. From the initial assignment in early 1971

to the last transfer all laboratories have shown changes of

less than 1 ppm per year. Neglecting any adjustments for

drift, the magnitude of the change in the units is only

0.28 nV per year with a 3 standard deviation uncertainty

of 0.53 mV. In addition, Phase I type of experiments, by

themselves, provide a powerful tool for evaluating a group

of laboratories simply and quickly. In fact, this approach

would be a very useful tool, when conducted between

national laboratories, to unravel the unresolved discrepancies in measuring the difference in as-maintained units of voltage reported at the 1972 Conference on Precision Electromagnetic Measurements by Eicke and Taylor [1]. Finally, the results to date demonstrate the feasibility of establishing regional volt maintenance programs. In fact such techniques can and are now being extended to areas such as resistance capacitance.

ACKNOWLEDGMENT

The results reported here and the success of the experiment were made possible by the dedicated efforts of a large

number of individuals at the participating laboratories. At

the risk of overlooking someone, we gratefully acknowledge

the collaboration of the following persons: B. Morton of Beckman Instruments Co., R. Wangerin and R. Stromquist of Electrical Standards Repair Service, D. Buck and

R. Englander of Lockeed-California Co., F. Woodsmall

and H. Buehrle of Rockwell International, and R. Stodola

and G. Davidson of TRW Systems Group. Special thanks

are due B. F. Field of NBS for coordinating the Phase I experiment, R. E. Kleimann and G. Free of NBS for co- ordinating the Phase II experiments. We are particularly

grateful to H. H. Ellis of NBS for the hundreds of measure- ments required on the transport standards. REFERENCES

[1] W. G. Eicke, Jr., and B. N. Taylor, "Summary of international comparisons of as-maintained units of voltage and values of 2e/h," IEEE Trans. Instrum Meas., vol. IM-21, pp. 316-319, Nov. 1972.

[2] "NBS offers new calibration service for the volt," Nat. Bur.

Stand., Washington, D. C, NBS Special Publ. 250, Users Bull,

no. 1, Mar. 1971.

[3] W. G. Eicke and J. M. Cameron, "Designs for the surveillance

of small groups of saturated standard cells," Nat. Bur. Stand., Washington, D. C., Tech. Note 430, 1968.

[4] , "NBS test reports for the volt transfer program," unpublished.

[5] B. F. Field, T. F. Finnegan, and J. Toots, "Volt maintenance

at NBS via 2e/h: A new definition of the NBS volt," Metrologia,

vol. 9, no. 4, pp. 155-166, 1973.

Reprinted by permission from IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT

Vol. IM-23, No. 4, December 1974, pp. 290-294 Copyright 1975 by the Institute of Electrical and Electronics Engineers, Inc. PRINTED IN THE U.S.A.

331
A COORDINATED SYSTEM OF MAINTAINING AND DISSEMINATING THE VOLT,

Robert M. Shaw, IEEE Trans. Instrum. Meas. IM-29, December 1980.

Groups of standard cells may be used to maintain the volt with an uncertainty of 1/4 parts per million (ppm) using a separate transportable

group of cells to transfer the value from the National Bureau of Standards

(NBS). An enclosure to hold 24 standard cells has been designed in which

temperature control is better than 100 yK in an environment varying as much

as 3 K. The volt thus maintained is transferred over coaxial cables to

service groups about 1/2 km away. This is done by generating a constant

current that produces a 1-V drop across standard resistors located at both

ends of each cable group. Accuracy of the transferred volt is about 1 ppm.

THE RESPONSE OF STANDARD CELLS TO ALTERNATING CURRENTS IN THE FREQUENCY RANGE

10-105 Hz, G. J. Sloggett (National Measurement Laboratory, CSIRO, University

Grounds, City Road, Chippendale, Australia 2008), Metrologia j_3_, 57 _

61

(1977).

Measurement of the effects of passing alternating currents in the frequency range 1 0-1 0^ Hz through a group of saturated standard cells showed

that their internal resistances declined with increasing frequency at a mean

rate of 25 fl/decade. AC caused the emf's of certain cells to decrease,

whereas only increases have been previously reported. For most cells, this

effect became very small for frequencies above 1 kHz. Also, a previously

unreported phenomenon, the generation of an emf at twice the ac frequency,

has been observed. A simple relationship between the magnitude of this

effect and that of emf change is suggested.

AUTOMATIC INTERCOMPARISON OF STANDARD CELLS, Andrew F. Dunn, IEEE Trans.

Instrum. Meas. IM-23 , December 1 974

A system is described whereby precise intercomparisons of standard cells

are accomplished under full computer control. The results obtained indicate

accuracies (one standard deviation) of ±10 nV from which the individual

values of the standard cells in a maintaining group of ten cells are estimated to be accurate to ±20 nV. The major source of indeterminacy continues

to be the variation of standard cell EMF due to temperature, mechanical

vibration, etc.

332
INTERNAL COMPARISON OF A LARGE GROUP OF STANDARD CELLS, J. Wilbur-Ham, IEEE

Trans. Instrum. Meas. IM-22, September 1973-

Equipment is described for comparing any number N of standard cells

which constitute a standard of electromotive force. The scheme is

economical, needing only 2N measurements. The cells are disposed in a ring,

and there is simple switching to compare each cell with its four neighbors.

The calculations are done by a program on a desk calculator. They give the

value of each cell with respect to the mean of the group. They also give the rms error, which is typically less than 0.02 yV for one cell in a group of

22. There are safeguards for identifying a spurious measurement and a false

keyboard entry.

AUTOMATIC MEASURING SYSTEM FOR A CONTROL OF STANDARD CELLS, Hiroyuki Hirayama

and Yasushi Murayama, IEEE Trans. Instrum. Meas. IM-21 , November 1972.

The automatic measuring system developed in the Electrotechnical

Laboratory to monitor standard cells requiring a lot of measurements is

described. It is composed of a scanner, an integrating-type digital voltmeter, a programmer or minicomputer, etc., and carries out the data acquisition and processing for a maximum of 200 cells. The difference in EMF of two

cells is measured precisely. To reduce the effect of the induced EMF in the

scanner, a delay unit is provided, and procedures minimizing errors and

evaluating random errors have been adopted. In the on-line version of

system, a great part of the data processing is done during the delay and

integrating time of digital voltmeter. Applying this system to the

measurement of standard cells, a precision of the order of 0.1 yV has been

obtained.

MAINTENANCE OF A LABORATORY UNIT OF VOLTAGE, Andrew F. Dunn, IEEE Trans.

Instrum. Meas. IM-20 , February 1971.

Many of the problems involved in maintaining a laboratory reference

group of standard cells with a reliability better than one part per million

are outlined and solutions suggested. Methods are presented for reducing the

number of measurements by approximately M0 percent from the ideal without

seriously reducing the accuracy achieved, and procedures for assessing the

quality of the group are indicated. Suggestions are given for preferred

methods to be used for groups of 4, 8, or 12 cells, and as an example of

these procedures the 20-month history of a new group of standard cells is

illustrated.

333
TRULY TRANSPORTABLE STANDARD-CELL AIR BATH, David W. Braudaway, IEEE Trans.

Instrum. Meas. IM-1 9, November 1970.

Dissemination of the volt is complicated by the complex temperature

response of standard cells. Described are the design and performance characteristics of an experimental standard-cell air-bath specifically optimized

for transportable voltage-dissemination service. Salient features of this

bath are 1) ±0.001 °C control of the cell temperature in the laboratory,

±0.02°C maximum variation around 28°C in shipment; 2) exposure capacity from

over 8 hours at 0°C or H5°C to over 72 hours in moderate temperatures;

3) shipping package weight 14.5 kg, size 20.3 x 22.9 x 30.5 cm; and

4) electrical power drain 0.8-0.2 watt.

Cell temperature is determined primarily by an ambient-temperaturebiased on-off heater system. Immunity to external environment is achieved by

a unique secondary control, which employs the energy storage capacity of two

heat-of-fusion alloys, which are reset simply by exposure to laboratory

temperature

Experience with this bath has shown that the average value of cell

voltage is reproducible to 0.1 pV within a few hours following shipment.

This time delay is significantly shorter than the days-to-weeks for most air

baths and the weeks-to-months for cells shipped without temperature control.

THE APPLICATION OF THE DIRECT CURRENT COMPARATOR TO A SEVEN-DECADE

POTENTIOMETER, Malcolm P. MacMartin and Norbert L. Kusters, IEEE Trans.

Instrum. Meas. IM-1

, December 1968.

The design and construction of a self-balancing direct current

comparator for use in a seven-decade potentiometer is described. The

comparator generates an output current whose value, as a proportion of a constant input current, is determined to a very high accuracy by the ratio of

the numbers of turns of two windings on a magnetic core. A linear,

adjustable voltage scale is obtained by passing this output current through a resistor whose value does not vary with current. Since the voltage adjustment is made by varying turns on a magnetic core and not by means of a resistive divider, the usual problems of contact resistance and thermal

electromotive forces associated with this adjustment in conventional

potentiometers are avoided.

The main sources of error in the comparator and the design techniques

used to keep the errors less than the smallest step of the output current is

discussed. A self-checking feature whereby the linearity of each step of the

output current can be checked quickly and easily is described. The

performance of the prototype model is given. The normal range of the

potentiometer is from zero to 2 volts in steps of 0.1 yV.

334
THE CONSTRUCTION AND CHARACTERISTICS OF STANDARD CELLS, George D. Vincent

(Director, Standard Cell Dept., The Eppley Laboratory, Inc., Newport, RI) IRE

Transactions on Instrumentation 1-7, 221-23^ (1958).

In order to maintain standards of the electrical units, two of the three

quantities interconnected by Ohm's Law must be represented by stable

reference standards. The best standard of EMF now known is the cadmium

standard cell. The saturated cadmium standard cell, carefully made and

maintained, as a reproducibility and stability in the order of one or two

ppm, which is about one order of magnitude better than the accuracy at

present attainable in the measurement of the absolute dimensions of the

electrical units. Probably in the not-too-distant future, refinement in the

determination of the electrical units in terms of fundamental standards, or of a natural constant, will require even more stable standards of reference.

335

RESISTANCE AND RESISTANCE APPARATUS

Automated NBS Measurement System

Kenneth R. Baker and Ronald F. Dziuba (1983) 339

A New Switching Technique for Binary Resistive

Dividers

Robert D. Cutkosky (1978) 344

NBS Automated One-Ohm Resistance Measurements

Jerome J. Morrow (1976) 346

An Integrated System for the Precise Calibration of

Four-Terminal Standard Resistors

Thomas E. Wells and Earl F. Gard (1971 ) 350

Resistive Voltage-Ratio Standard and Measuring Circuit

Ronald J. Dziuba and Bernadine L. Dunfee (1970) 355

Evaluation of the NBS Unit of Resistance Based on

a Computable Capacitor

Robert D. Cutkosky (1961) 367

Alloys for Precision Resistors

C. Peterson (1954) 379

The Error Due to the Peltier Effect in DirectCurrent Measurements of Resistance

C. G. M. Kirby and M. J. Laubitz (Abstract, 1 973) 393

Resistance Comparisons at Nanovolt Levels Using

an Isolating Current Ratio Generator

L. Crovini and C. G. M. Kirby (Abstract, 1 970) 393

337

IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-32, NO. 1, MARCH 1983

Automated NBS \-Q Measurement System

KENNETH R. BAKER, STUDENT MEMBER, IEEE, AND RONALD F. DZIUBA, MEMBER, IEEE

Abstract—A microcomputer-controlled measurement system has

been developed for calibrating stable, 1-ft standard resistors. It consists of a direct current comparator potentiometer, a serf-balancing detector

circuit, and special switching networks. The measurement system is capable of comparing resistors to a precision of better than 0.01 parts per million (ppm).

I. Introduction

THE UNIT of resistance is maintained at the National Bureau of Standards by a group of five Thomas-type 1 -Q

resistors. A dc current comparator potentiometer is used to compare the NBS reference group to other resistors under

test [1]. Fifteen resistors, five of which comprise the reference group along with two control resistors and eight unknown re- sistors, can be connected in series with the primary circuit of

the dc current comparator. The value of any resistor in the

string can be determined by indirectly comparing its voltage

Manuscript received September 8, 1982; revised October 29, 1982.

K. R. Baker is at the National Bureau of Standards with the Co-operative

Engineering Education Program, University of Maryland, College Park, MD

20742.

R. F. Dziuba is with the Electrical Measurements and Standards Division, National Bureau of Standards, Washington, DC 20234.

drop to the mean voltage drop of the reference group via a

stable "dummy" resistor in the secondary circuit of the comparator. The resistors in the primary circuit are measured at a current level of 0.1 A (0.01 W/resistor); therefore, a detector

resolution of 1 nV is required in order to measure a resistance change of 0.01 parts per million (ppm). The Thomas-type

resistors are hermetically sealed in double-walled enclosures and are very stable after being fully annealed; however, be- cause they are constructed of manganin they exhibit significant

temperature and pressure coefficients. Thus to achieve ac- curacies of the order of 0.01 ppm, the operator must monitor

the bath temperature of the resistors along with the ambient

conditions, primarily the barometric pressure. The operation of the system in the manual mode is limited to a resolution of 0.025 ppm, which corresponds to an interpolation of one-half

of the last dial setting. Automation of the system has been a high priority not only because of the large work load in the calibration of 1-12 standards and the ease with which a computer can monitor the significant test parameters, but also because of the possibility of improving the resolution and ac- curacy of the system.

An earlier attempt was made to automate the system by

using a digital nanovoltmeter as a detector in the critical part

339
of the dc current comparator circuit [2]. This detector could be readily interfaced to a computer; however, it proved unsatisfactory due to excessive ac output from the comparator

which introduced noise greater than 50 nV rms in the detector

circuit. The present automated system has alleviated these problems and has achieved a resolution of better than 0.01 ppm

in the comparison of stable resistors. At this resolution, the practical measurement range of the automated system is ±500 ppm; however, it can easily be extended to wider ranges

at slightly reduced accuracies. Over 95 percent of the Thomas-type resistors measured at NBS have deviations within ±50 ppm from 1 fi. A few resistors constructed prior

to 1 948, whose values were based on the "international" units, have deviations at the ±500-ppm level. Thus the ±500-ppm

range of the automated system satisfies the measurement needs

of NBS in this area.

DEMODULATOR

Fig. 1 . Schematic diagram of dc comparator potentiometer circuit.

II. System Description

A. Comparator

The heart of the measurement system is the dc current comparator as shown in Fig. 1 . In the dc current comparator,

a condition of magnetic balance in the core is detected by

modulating the permeability of the core by a square-wave

modulation current. The output of the demodulator is pro- portional to the magnitude and polarity of the ampere-turn

imbalance and it is used to correct the slave current source to obtain a zero flux condition in the core [3]. The comparator

is a commercially available unit having an adjustable 2000-

turn primary, 1000-turn secondary, and a fixed 0.1 -A pri- mary current source. In the automatic mode, the primary

winding is set at 2000 turns and a relay with mercury-wetted

contacts is used to insert a unit winding as a 500-ppm cali- bration signal.

B. Resistors The 1-Q resistors in the primary circuit are connected in series on a piercury ring stand which is totally immersed in a specially designed circular oil bath with temperature maintained at 25.000 ± 0.003°C. A modified crossbar switch selects the appropriate potential terminals of a resistor in the primary

circuit whose voltage drop is to be compared to the voltage drop

of the 0.5-ft resistor in the secondary circuit. The 0.5-17 resistor or "dummy" is a specially designed resistor having a low load

coefficient and a negligible drift during the period of a test run.

C. Detector and Feedback Circuit The voltage difference Vx between the dummy and test resistor is detected by a photocell-galvanometer amplifier

(PGA). The PGA has been modified by mounting a pair of cadmium-sulfide (CdS) photoconductive cells adjacent to the

original set of photocells in order to provide an isolated output

[4] . The CdS cells form two arms of a Wheatstone bridge, as shown in Fig. 2. The output of this bridge yields a sensitivity of 0.067 V/ppm. This output voltage is buffered by an operational amplifier, A \ and then amplified by a factor of 5 by AjSince a 1-ppm change of resistance corresponds to a voltage

change of 0. 1 j/V, the open-loop gain of the system G at this

point is equal to 3.35 X 106. It was observed that any further amplification will lead to oscillations in the feedback circuit. Amplifier /I3 provides a feedback current // through the 10- turn winding A7/. This 10-turn winding is normally used to provide adjustable standardization settings to make the dc

current comparator potentiometer direct reading. The current

// is monitored by measuring the voltage drop across a 1 00-12

resistor with a digital voltmeter (DVM). A 1-mF capacitor of PTFE dielectric connected across the 100-Q resistor provides

sufficient filtering of electrical noise in the system. The feed- back system is similar to the one described in [5]. At zero flux in the core

and at detector balance V/

RrI

n - RJS

(1)

(2) The value of resistor Rr can then be expressed as

Nph.

The difference in value between resistors 1 and 2 in the primary

circuit can be expressed in terms of the difference in their re- spective feedback currents at balance, i.e.,

Since the feedback current difference can be expressed as the voltage difference A V across the feedback resistor Rj then

R< Nr AV

P R/N.s IP and

AV _NsR[

RrIp

(3)

(4) (ARP/Rr ) Nf Rs which gives a sensitivity of 20 ^V/0.01 ppm.

Amplifier Aj is capable of supplying ±10 mA which would

correspond to a measurement range of ±500 ppm using this feedback circuit. A booster amplifier could be used to increase

the measurement range of the system.

It is essential to examine whether the system has sufficient

340
G = 3.35X10 8 R3 < 6" I.J-^IO turns

V3 = GV. J_

l,= V3 /R3 V

Fig. 2. Schematic diagram of detector and feedback circuits.

gain to provide the necessary feedback control for large re- sistance changes. As shown in Fig. 2, the feedback current can

be expressed as GVx

(5)

where R] is the controlling element of current amplifier Ay.

A change in Rp

is reflected as a change in Vx , i.e.,

&Rn = (6)

Substituting (1) and (5) into (6) and then dividing by Rn gives

ARP Np Rj

Rn

AIS Nf GRP

Is (7)

where Als/ls represents the normalized measurement range and ARp/Rp

indicates an apparent offset caused by insufficient gain in the feedback system.

The offset for a measurement range of 1 50 ppm is 5 X 10-8

, calculated from the values given in Figs. 1 and 2. This offset can be seen as a deflection on a secondary galvanometer that

is connected to the nonisolated output of the PGA. It has been

observed experimentally that a 150-ppm change results in a 1-mm deflection on the secondary galvanometer. This deflection corresponds to an offset of 4 X 10-8 which is in rea- sonably good agreement with the calculated value. For a measurement range of 2800 ppm, the offset Would be 1 ppm

which corresponds to a 25-mm deflection on the secondary

galvanometer.

The offset problem can be eliminated if the change of current in the feedback circuit is measured for a known change

in ratio. This is accomplished by measuring the change in voltage across the 100-1] resistor in the feedback circuit before and after inserting a unit winding in the primary circuit. For

a 2000-turn primary, this corresponds to a 500-ppm change

in ratio. Then the offset will not result in a measurement error

if the detector output is linear and the system gain is constant. Measurements indicate that these factors contribute less than

a 5-percent error for an offset of 1 ppm or less. Thus for measurements of 1-fl resistors over the range of 0 ± 500 ppm, the

resulting error due to insufficient gain in the feedback circuit

would be less than 0.01 ppm. The error would increase to 0.05 ppm for resistors that deviate from nominal by 2800 ppm.

D. Microcomputer

The microcomputer is a disk-based system with an 8-bit microprocessor and 64K of RAM. It utilizes the S-100 bus and

comprises eight parallel input/output ports, an IEEE-488

controller port, a 1 2-bit A/D converter port, two RS-232 ports, and a clock board. Basic I/O is accomplished with a CRT

terminal and a printer via the RS-232 ports. The microcomputer is completely dedicated to running the measurement

system. A block diagram of how the computer interacts with

the other parts of the system is shown in Fig. 3. Interfacing of the resistor selection switch, detector sensitivity switch, ambient temperature and humidity sensors, as

well as the barometric pressure transducer was done via the eight parallel I/O ports. Ambient temperature and humidity

are monotired by a digital thermometer/hygrometer instru- ment that is interfaced to the computer through two of the I/O

ports. The barometric pressure is monitored with a transducer whose frequency output is proportional to the input pressure. Transducer temperature is determined indirectly by measuring

the voltage drop across a calibrated diode that is mechanically

attached to the tranducer body. The pressure sensitivity of the system is 0. 1 mmHg ( 1 3.3 Pa) and the corresponding accuracy

is checked periodically against a calibrated aneroid barometer. The pressure transducer has been tracking the aneroid ba- rometer to within 0.3 mmHg which is sufficient accuracy for applying pressure corrections to 1-fl resistors. The transducer outputs are measured by a timer/counter/DVM that is controlled by the microcomputer through a single port. The DVM,

which measures the feedback current, communicates along

the 488 bus. A platinum resistance thermometer (PRT) bridge monitors the oil bath temperature. The recorder output of this bridge is interfaced to the microcomputer by the 12-bit A/D

converter. This approach provides a temperature measurement

range of 25.000 ± 0.010°C with a resolution of 0.1 mK.

E. Critical Switching

The critical switching area in this automated measuring

system is that necessary to select and connect the potential

circuits of the resistors to the PGA detector. Although the

341
AMBIENT f TEMPERATURE >

AMBIENT

HUMIDITY

BATH TEMPERATURE

BAROMETRIC

PRESSURE

DIRECT CURRENT COMPARATOR

-500 ppm OFFSET CURRENT REVERSAL

D.AL SETTING>

PE

SE ETECTOR

NSITIVITY \j

RESISTOR

SELECTION SWITCH

Fig. 3. Block diagram of microcomputer-controlled measurement

system.

reversal technique used for achieving a balance condition

cancels the effect of constant thermoelectric voltages generated

in switching this part of the circuit, it is desirable to minimize

the magnitude of these voltages in order to reduce the effects of varying thermal EMF's. A proper sequence of data taking can cancel the effects of thermal EMF's that are varying lin- early.

The resistor selection is of crossbar type having low thermal

contacts and is similar to the one described in [6]. The acti- vating coils of the switch are separated from the switch mechanism to reduce heat transfer to this critical area. The

switch mechanism is housed in a heavy aluminum box to suppress thermal gradients.

The detector sensitivity switch consists of four latching-type low thermal relays. The relays are sealed in a heavy aluminum

box filled with silicone fluid and the box is immersed in the oil bath that contains the resistors. When operating in the manual

mode, all four sensitivity positions are needed to adjust the primary turns of the comparator for a balance condition. In the automatic mode, only the highest sensitivity relay is re- quired since the feedback circuit protects the PGA detector by servoing it to a null condition. Even if the potential terminals were connected mistakenly in the wrong polarity the offset voltage at the PGA would not be enough to do permanent

damage.

F. Software

The computer program is written in BASIC language using

multiple subroutines to handle the data taking and data processing. The flowchart shown in Fig. 4 indicates the specific

operations performed by the microcomputer.

Q

START ^

Clear

Syster,

Final Calculations

(

E "° ) Ambient Temp. a Hu

KTa« Sensitivity Walt 25 se<

Negative I Direction! Read DVM

Walt

1 2 aec

I Positive Direction

600 ppm

offset In

Error Routine

J Positive Direction

Reverse

Current

Negative

Direction

600 ppm

offset out

Sensitivity Out

Resistor Out

Calculation

Fig. 4. Flowchart of the routine for \-Q measurements.

III. System Performance

The data in Table I are given to show the level of precision

that is available with the automatic system. The same set of

resistors were measured four times within a single day and the

differences from the mean of these four series of measurements

for individual resistors are listed in the table. The number of DVM readings that were taken and averaged per resistor was

varied from 8 to 20 readings. The table lists the DVM readings

according to the actual time sequence in which the data were

taken. When a resistor was measured one-half of the total DVM readings was taken for one particular current direction, then the current direction was reversed, and after an approximate 1 2-s delay, the remaining half of the DVM readings was

taken. The integration time per reading was 10 s. The larger

imprecision for data using the 20 DVM readings is probably caused by varying thermal EMF's. No significant improve- ment is obtained if several reversals are performed for each

resistor. The measurement range was <40 ppm.

The data in Table II indicate the differences in parts per

million between the automatic and manual measurement

modes. For the manual mode, an operator varies the primary

turns to achieve a null balance with the PGA detector. Four measurements are taken on each resistor in the manual mode

with a resolution of 0.025 ppm for each measurement. Since

the automatic system is more precise, the differences are mainly due to the lower precision and operator bias in the

342
TABLE I Precision of Automatic System

(Differences from mean value in parts per million.) MEAN NUMBER OF DVM READINGS

RESISTOR VALUE 1

2

10 8 20

C93 33.1908 - 001 -.001 -.00

1

003 S66 18.4700 -.001 -.002 .002 001 S69 3 0070 .001 -.004 -.001 .004 S70 -3.9908 -.003 -.000 -.002 .005 S72 1.2134 -.003 -.002 -.001 006 S83 17.0494 -.001 -.003 -.005 .009 C84 6.1205 -.004 .002 -.005 008 #8 34.0921 -.002 .001 -.002 .004 #9 28.0882 -.001 .002 -.002 001

10 33.1591 - 001 .001 -.001 .000 #11 33.57 14 .003 -.000 -.004 001 #12 33.4628 -.004 .007 -.002 - 002

13 2.1992 -.003 -.004 -.002 .009 #14 35.4442 -.002 .004 .000 -.002 #15 32.4392 001 002 -.002 - 001 mean -.0014 0002 -.0019 .0031

s.d. 0020 0030 0018 0038

TABLE II Differences Between Manual and Automatic Measurement

Modes

(Differences in parts per million.)

RESISTOR 5/28/82 6/01/82 6/03/82 6/07/82

C93 -.001 -.006 -.020 - 008 S66 .005 .013 01 1 003 S69 .013 -.017 -.014 -.005 S70 -.029 012 .013 .000 S72 018 015 .008 -.009 S83 - 007 -.022 -.018 011 C84 .040 .039 024 - 027 #8 -.010 -.040 -.018 -.029 #9 .003 007 .01 1 -.002 #10 .01 1 016 .025 -.005 # 1 1 .018 .025 012 - 047 #12 .025 .033 .028 023

13 .020 .027 .020 007 #14 .019 .028 -.000 .020 #15 .026 .029 -.005 007

mean .010 .01 1 .005 -.004

s.d .017 023 .017 .019

manual mode. The random nature of the differences indicates no measurable systematic error between the two measurement

modes.

IV. Future Plans

Future plans include an evaluation of systematic error of

the automatic system and then a re-evaluation of the uncertainty [0.08 ppm (3o-)] that is presently assigned to l-fi measurements at NBS [1]. The automated system has improved the precision in comparing \ -Q resistors. With an in- crease in the number of measurements and the improved

monitoring of test parameters that the automated system

provides, it should be possible to obtain a better estimate of the

possible systematic errors in the measurement process. It is planned to have the microcomputer transfer the data to a

time-shared minicomputer where the data will be permanently

stored and a more complex data analysis performed.

With this system it is possible to measure resistors that are

located in a separate oil bath with no apparent degradation of

precision or accuracy. Measurements have been made successfully on resistors in a variable-temperature oil bath with

current and potential leads over 7 m long. Thus it is possible

to compare resistors at any temperature against the NBS

reference group maintained at 25°C. This capability is important because many international laboratories maintain their standards at 20°C and consequently it is desirable to compare

standards at their working temperature. Unfortunately, the present thermometer bridge does not have the capability for computer control over a wide range of temperatures. An au- tomatic resistance thermometer bridge [7], which can be

readily interfaced to the microcomputer, is being built to measure PRT resistances over the range 0-100 Q. Another

feature of this capability is that it now becomes possible to automatically determine the temperature coefficients of re- sistance for standard resistors.

V. Conclusion

The automatic system has improved the precision of comparing 1-fi standard resistors. It is expected that, after a re- evaluation of the uncertainties in the measurement process, an improvement of the accuracy level of these measurements

will be demonstrated. Another advantage is that the automatic

system relieves the operator from tedious repetitive measurements using a light-beam galvanometer.

Acknowledgment

The authors wish to thank R. Pederslie and D. Prather for assisting in the experimental work. The authors are also grateful to T. M. Sounders for helpful discussions on current comparators and to B. Field for suggestions on analog and

digital circuits for the system. Last, but not least, the authors wish to thank R. Fronk for his assistance is solving their soft- ware problems and for his continued help in developing the necessary software required to interface the microcomputer

with the minicomputer.

References

[ I ] T. E. Wells and E. F. Gard, "An integrated system for the precise cali- bration of four-terminal standard resistors," IEEE Trans. Instrum. Meas.,vo\. IM-20, pp. 253-257, Nov. 1971.

[2] J. J. Morrow, "NBS automated one-ohm resistance measurements,"

Quality Progress, pp. 22-25, Nov. 1976.

[3] P. McMartin and N. L. Kusters, "The application of the direct current comparator to a seven-decade potentiometer," IEEE Trans. Instrum. Meas., vol. IM-17, pp. 263-268, Dec. 1968.

[4] T. F. Finnegan, "AC Josephson effect determination of e/h: A standard

of electrochemical potential based on macroscopic phase coherence in superconductors," B.S. thesis, Univ. of Pennsylvania, 1971.

[5] H. van Kemper, H. W. Heyenhuisen, and J. H. J. M. Ribot, "Semiautomatic bridge for high-precision dc resistance measurements on pure

metals at low temperatures," Rev, Sci. Instrum., vol. 50, pp. 161-164,

Feb. 1979.

[6] D. W. Braudaway and R. E. Kleimann, "A high-resolution prototype system for automatic measurement of standard cell voltage," IEEE

Trans. Instrum. Meas., vol. 1M-23, pp. 282-286, Dec. 1974.

[7] R. D. Cutkosky, "An automated resistance thermometer bridge," IEEE

Trans. Instrum. Meas., vol. IM-29, pp. 330-333, Dec 1980.

343
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-27, NO. 4, DECEMBER 1978

A New Switching Technique for

Binary Resistive Dividers

ROBERT D. CUTKOSKY, fellow, ieee

Abstract—A binary resistive divider network has been found which

utilizes N cascaded DPDT reversing switches plus one SPDT switch

for an V-bit divider. The network features a simple and accurate method for internal calibration. With a fixed divider input voltage the power dissipated in each resistive element is independent of switch

setting.

I. Introduction

MANY OTHERWISE satisfactory resistive dividers

are found in practice to be very difficult or time

consuming to calibrate. This problem is most effectively

handled during the design of the instrument, at which time

much future frustration can often be eliminated. The resist- ive divider described here contains its own internal calibra- tion circuitry, but it would probably be an impractical device if it were not coupled to a microprocessor. It is hoped

that the technique contains enough advantages to make this additional complexity worthwhile.

II. Basic Circuit

The network shown in Fig. 1 is proposed for use as a binary resistive divider. It can be seen that with no output

load, the resistance of the network to the right of each DPDT reversing switch is always 2 (arbitrary) units of

resistance, so that with Kjn fixed, the power in each resistance element is independent of all switch settings. It also can be

seen that if all switches are on "0" (as represented in Fig. 1 ), Koul = 0; if Si is on "1" and all other switches are on "0",

K>ui = ^n- All intermediate output voltages representable by

Manuscript received June 5, 1978: revised July 27, 1978. The author is with the FJectrical Standards Division. National Bureau

of Standards. Washington, DC 20234.

Fig. 1. An /V-bit binary resistive divider Relative resistances are

an N-bit binary code can be obtained by two different settings of the cascaded switches. This redundancy can be

used to provide the divider with a self-calibration facility. Consider the network with all switches on "0" except SN + , (switch code 0—001). Then, Kout /Kin = 1/2", If the switch code is now changed to 0—011, one again has VouJVin =

1/2". The effect of the operating switch N is to interchange

the dropping resistor and the load resistor to the right of S N.

If Vou{ is observed to change, the difference can be retained as a correction for the Nth divider. In a similar way, one cart measure the change in Vout when

the switch code changes from 0—0010—0 ((k + l)th switch

set) to 0—0110—0 (kth and (k + l)th switch set) to obtain

the correction to the kxh divider. Note that VoM /Vin = l/2

k and that the load resistor for the kth divider contains all of the resistance elements to the right of stage k\ since this resistance is independent of all switch settings, however, no

error will occur when the individual corrections are combined. It is therefore sufficient to make N difference measurements (2N individual measurements) to completely

calibrate the binary divider. The N difference measurements required in the above

scheme could be performed with ease if a fixed voltage

344
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-27, NO. 4, DECEMBER 1978

Fig. 2. An N-bit divider with enhanced self-calibration capability. All resistances are nominally equal.

source were available which provided constant output

voltages programmable to approximate each of the N

voltages to be compared. An ordinary N-bit uncalibrated

divider could be used for this purpose provided it had the

requisite stability. Several ways can be found for generating the required

reference voltages within the divider itself, one of which is shown in Fig. 2. To calibrate the kth divider, first set switches

k - 1 and k + 1 to "1", and set all other switches to "0". Measure the ideally zero voltage, Koul - Vitsl . Second, set switch k to "1", leaving the other switches as they were. Any

change in Voul - Ktes, can be attributed to an error in the klh

divider.

III. Ratio Calculations

In the following analysis, let Sk be the logical value of the kth switch; that is, Sk = 0 if the kth switch is "down" as shown in the figures, and Sk = 1 if it is reversed. Using the nomenclature of Fig. 3, neglecting switch and wiring resist- ances and writing ek = (Zu — ZDk)/(ZLk + Z^), one has V1 /Vin = S1 , KW^l-Sj.and

v"jv.m = 1/2 + (i -asifc/i

More generally, V'k is obtained by replacing Sk by 1 — Sk in the expression for Kk ,and Vk =(V'k + Vk )/2 + {V'k - Vk)eJ2.

The recursion formula

WK„= K/V,n + (Sk + i /2

k

) fl (l-2Sp)(l+ep)

can be readily verified, from which it follows that VN+l /Vin = S 1+ S2 (l - 2Sx)(l + et )/2 + S3 (l - 2S2 )

• (1 - 2S X )(1 + £l )(l + £ 2 )/4 + S4(l ~'2S3 ) (l-2S2)(l-2S1 )(l+e,)(l+e2) .(i + «a y8 + -.

For calibration of the fcth ratio, measure VN+1 /Vin = Voul /Vin first with Sk = 0, Sk+l = 1, and then

Fig. 3. The ~ih stage of a binary divider with voltages and impedances

labeled. The load resistance Zu is made up of all successive resistors in the chain. Zn is the dropping resistance.

with Sk = 1, St+1 = 1, in each case with all Sp = 0 for p > k + 1. The difference between these ratios is given by

for 2 < k < N, and A, = ej. Note, however, that the sign to be attached to the At

's must be reversed if during calibration an odd number of the switches to the left of the kth switch are

logically true. Ignoring terms of the order e

2 and higher, we have finally Kw+1 /K„*S1 +S2(l-2S1 )(l + A1 )/2 + S3 (l-2S2 ) (l-2S 1 )(l+A 1 +2A2 )/4 + S4(1-2S3 )(1-2S2 )

• (1 - 2S,)(1 + A, + 2A2 + 4A3 )/8 + ••

If each of the voltage measurements involved in the

calibration of the divider has an uncertainty of a h then each At will be uncertain by cr, v

/2. The total uncertainty in the corrected value for VN + l , <j t, may be calculated from

the preceeding equation. It is found that for an N-bit divider, the maximum uncertainty in VN+ ,,max(<r r), occurs when all of the switches are up (Sk = 1 for all k). To a very close approximation, max(<r T) % oiy/2yjN + 1/3. Although

these assertions about max(a T) are made wi thout proof, they

may be readily verified for realistic values of N.

IV. Practical Considerations

Resistive dividers achieve their highest accuracy when the

resistances are chosen so that the uncertainties due to switch

resistance variations are about equal to those due to leakage

resistance variations. If conventional switches or relays are

used, lOOOOO-Q resistors are about optimum and should

yield divider accuracies around 1 part in 10 7

. It is expected

that a commercial solid-state digital-to-analog converter would be used in the low-order end of a practical divider to reduce the number of relays to a minimum. A 17-bit DAC

combined with an 8-bit binary divider is being considered

for voltage measurements up to 20 V.

345
NBS Automated One-Ohm

Resistance Measurements

* Jerome J. Morrow

Physicist, Electricity Division

U.S. National Bureau of Standards

Automation of the U.S. National Bureau

of Standards' one-ohm resistance meas- urement system is part of a program of automating all systems used in calibration

of electrical reference standards. Auto- mating these systems allows more meas- urement redundancy in both number and

type of test, and will provide the opportunity for the NBS staff to gain experience

in automated systems which are rapidly

replacing manual systems in all parts of the measurement world.

In automating the NBS one-ohm manual

system it was necessary to preserve the manual system and to minimize the need

for new analog and digital hardware.

While these limitations are rather severe,

this approach offers several distinct ad- vantages. The automation activity may be

pursued with minimum interruption to on- going calibration work. Considerable sav- ing in captial equipment money may be

realized by utilizing the manual equipment to the largest extent possible. Finally, parallel construction allows the evaluation of critical components, such as switching, to be done by direct substitution.

•Contribution of the National Bureau of Standards. Not subject to copyright.

The manual system shown in Figure 1 consists basically of a dc current comparator potentiometer with resistors and

their switches in an oil-bath whose temperature is stable to better than ±0.001

degree Celsius. The following is based on

a simplistic view of the operation of the current comparator. Manual measurements consist of balancing the IR drops

across the "dummy" resistor in the secon- dary circuit and those of the unknown and

standard resistors in the primary circuit by changing the primary number of turns. Each balance provides an equation of the form:

Ri = (Npi/NS)RD

where Rj = resistor in primary circuit Npj = primary number of turns

for Rj

Ns = secondary number of turns

(constant for all balances)

Rjj — dummy resistor Since Ns and Rr_) are constant and five of the resistors are standards, it is possible

to determine the unknown resistors in terms of the mean of the five standards.

In order to automate and preserve such

a system, it was decided to place a modified cross-bar switch in parallel with the

existing manual rotary switch (for the primary resistors) and to add a switch to

346
the galvanometer circuit so that the

photo-cell galvanometer could be replaced

by a digital nanovolt meter (DNM). The

system would then be operated in an un- balanced state, i.e. the primary turns are not changed and the DNM would measure

the difference in IR drops across the dummy resistor and the resistors in the pri- mary circuit. The stability of the cross-bar

switch's thermal emfs and of the DNM

had to be several nanovolts or less in order to compete with the manual system.

From previous work in switching stand- ard cells, it was known that some com- mercial independent-select cross-bar

switches could be made suitable by separ- ating the electro- magnetic coils from the switch mechanism. The switch mechanism

is placed inside a heavy aluminum box and

the coils outside. The push rods from coil to switch point are extended by plastic rods. This type of separation reduces heat

transfer from coil to switch point suffi- ciently, and the heavy aluminum box effec- tively eliminates thermal gradients within the box. In addition all lines entering the box are thermally coupled to the box by means of one mil Mylar. Since the cross-bar

is in parallel with the manual rotary

switch immersed in the oil bath, it is possible to compare the magnitudes and stabilities of their thermal emfs. By manually

switching in a resistor and noting the deflection and noise on the photo cell galvo- nometer for each direction of current, and

repeating this with the other switch, it was concluded that there were no signifi- cant differences between the two switches. In addition, measurements of the same resistors with both switches yielded no sig- nificant differences.

Automating the Manual System

The DNM is a commercial item capable

of resolving one nanovolt with two nanovolts peak-to-peak noise for the loads and

voltages (10 microvolts or less) of the sys- tem. In terms of one-ohm resistors, one

nanovolt equals 0.01 ppm. However, all noise must be kept to a very low level be- cause of the low normal mode rejection of 70 db and common mode of 180 db inher- ent in the DNM. It was found necessary to

isolate the system from the power lines, to use coaxial cables for all lines between the current comparator and oil bath and, as

far as was possible, connect all grounds to the input ground of the DNM. Without

these measures the DNM output varies by

hundreds of nanovolts. The presence of pump motors and temperature controllers are probably responsible for most of the

residual noise. It also has been apparent

that line voltage fluctuations are responsible for some of the noise. This noise is also present in the manual system and ap- pears to correlate with the noise in the automated system. At present the peak-topeak noise in the automated system is usually ten nanovolts or less, and it is possible to reduce the noise by taking nine or more readings and averaging it out if nine

or more readings are taken. Resistance measurements in the auto- mated system consist basically of measuring the difference of the IR drops between

the "dummy" resistor and the resistors in the primary circuit. The primary and secondary number of turns and their currents are constant throughout all measurements. These measurements lead to equations of the form:

ISRD — IpRi = vj

where I s = current in secondary circuit R[> = dummy resistor Ip = current in primary circuit Rj = resistor in primary circuit

vj = voltage measured by the DNM

It should be noted that in the equations

for an unbalanced system the current di- rections and polarity of the DNM must be

known. The voltage, vj, is half of the dif- ference of the voltages for both current

directions. This difference eliminates ther- mal emfs and the DNM "zero" provided

they are constant within the time required

for the voltage measurements.

If such an equation for an unknown and

a standard resistor are subtracted from

one another, the result is: Rs — Rx = < vx — vs) /Ip

In order to obtain Rx , Ip must be measured. This is accomplished by switching a shorten place of the "dummy" resistor. The DNM will then read the voltage

across the standard from which one can

calculate the primary current. It should be

noted that the primary current measured

need not be very accurate since it is di- vided into a relatively small quantity. Also

note that the errors which are constant for

all measurements will be reduced further by the subtraction of the two voltages.

If the drift rate of the DNM "zero" is con347
Record <rTape Loop -

Program

Answer-Back

Subprogram

Teletype

Printer/ Punch

Reader/

Keyboard

Midas

System

Stimuli

^—Measure

ments

Recorders

Figure 2. MIDAS and teletype configuration.

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5832109* 5832109'* 5832)09* 5832109*

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FIGURE 3: Sample MIDAS paper-type program.

stant, the resultant "offset" error will b< eliminated. Errors such as voltage meas

urements on different DNM ranges wouli

not be reduced adequately. The one-ohn

resistor values measured by this systen

differ from nominal by less than 50 ppm oi five microvolts. Therefore, the DNM rang*

need not be changed.

The settling time of the automated sys

tern was found to range from 45 to 5J seconds. The former is for a resistoi change and the latter is for a current re

versal. The settling time for the DNM foi these same voltage changes with essen

tially no input noise is about 30 seconds. The settling time for the manual system is no more than two seconds with the auto

matic switch. These large settling times are a real disadvantage of the automated

system. If one ppm resolution is specified,

then the settling time would be six seconds or less. Control of the automated system was

accomplished by a modular programable

digital interface system developed at NBS. This system has been given the acro- nym MIDAS. In general, MIDAS consists

of a crate and power supply with a controller and modules which directly interface any device with TTL levels. The program

source and data logger can be any device which transmits and receives standard 7- bit parallel USASCII code format.

In the system considered here a stand- ard Teletype served as this device. Con- figuration of the MIDAS is shown in Figure 2. The operating rate of the sys- tem is limited by the Teletype of ten char- acters per second. This limitation was oi no consequence except on recording DNM

output. The A/D conversion rate is about two readings per second and there are five digits per reading plus several command

characters. Consequently about every

other reading was recorded. If the DNM

output is "flip-flopping" as a result of noise, it is possible to record only one side

of the variations, or considerably more of one side than the other. In addition, re- cording time constitutes a considerable

portion of the total run time. Another

limitation of this configuration is that any program must be sequential. This elim- inates any possibility of change in the se- quence, decision making, or determining

variable parameters such as settling time

and noise during a run. Coupling this lim- itation with the fact that no program could be a tape loop because of the need to switch resistors in the primary circuit in and out, experimentation with the sytem

required a large number of very long pro- grams. In the very near future, the

MIDAS will be controlled by a minicomputer and such limitations will be eliminated. 348
Differences between duplicate measurements in ppm.

Resistor

sition A/11/75 4/15/75 4/15/75 4/18/75 5/1/75 5/14/75 5/14/75

1 0.00 0.02 0 .01 -0.02 -0.01 0.02 -0.09

2 0.05 0.01 O 0 -0.02 0.27 -0.02 -0.08

3 0.08 0.01 -0 .02 -0.03 -0.04 -0.04 -0.04

4 -0.02 0.00 -0 .03 -0.02 0.04 -0.01 -0.03

5 0.02 0.03 0 .02 0.03 -0.03 -0.03 -0.02

6 0.00 0.04 0 .04 -0.02 0.00 -0.03 -0.01

7 0.00 -0.01 -0 .01 -0.01 0.02 0.02 0.00

8 -0.01 0.01 -0 .01 0.00 0.01 0.00 0.05

9 -0.01 0.07 -0,.03 0.00 -0.01 0.03 -0.01

10 0.00 0.01 -0..02 -0.01 -0.08 -0.02 -0.03

11 0.00 0.00 -0.,04 0.00 -0.06 0.00 -0.05

12 0.00 0.02 -0. 01 0.00 -0.09 -0.04 -0.05

13 0.00 0.01 -0. 01 0.01 -0.07 0.00 -0.02

14 0.00 0.00 -0. 02 -0.02 -0.07 -0.03 -0.03

15 0.00 0.04 0. 01 0.00 -0.07 0.02 -0-01

A portion of a tape program is shown in Figure 3. The basic sequence for the meas- urements for one resistor is, in this case, a

set of six lines in the figure. In terms of Teletype lines, it is a set of two. The first line prints the current direction (R); in- hibits the MIDAS for 60 seconds to allow

for settling of all components; and then

prints overflow, range, and three DNM

readings. Each reading is printed after a conversion pulse is received from the

DNM. The second and third lines print six more DNM readings. The fourth line prints the new current direction, the

current reversal relay is actuated and the remainder is the same as the first line. The fifth and sixth lines are identical to the second and third. The seventh line is the same as the first except that the old resistor is switched out and a new one

switched in. Evaluation

The stability of the automated system

has been measured by making duplicate measurements. As seen in Figure 4, there are only two differences greater than

0.1 ppm. Each day's measurements were made in the following order: 1 through 15 and then 15 through 1. Therefore the time

lag between both measurements of resis- tor 1 can be quite long. Depending upon the program, it ranged from 1.5 to 1.1 hours.

In the manual system a three sigma dif- ference of 0.056 ppm or less is acceptable.

In the automated system some differences larger than this are'simply due to noise. In other cases, it is due to a shift in the measured voltages. These shifts can be detected by finding the change in the measured voltages for each current direction

for both measurements of each resistor. These changes indicate that there is generally a very slow drift with time, usually

less than a nanovolt from one resistor to another. However, sudden erratic changes

of ten nanovolts or more occur. If they

occur between the voltage measurements

for both current directions, they will not be subtracted out. It has not been possible

to ascertain the cause of the shifts. Ther- mal emf shifts can be ruled out. The accuracy of the automated system

can be evaluated by comparing its measured resistance values with those from the manual system. With one exception, dif- ferences between resistor values from the automated and manual systems are 0.1 ppm or less. A least squares fit of a straight line to the nominal value corrections for 59 re- sistors from both systems yielded the

following line. Ca = -0.008 + 1.00013 Cm

where Ca = correction in ppm as meas- ured on the automated

system

Cm = correction in ppm as measured on the manual

system

The residual standard deviation of the

fit was 0.023 ppm. If both systems gave

identical results, then the equation would

be

Ca = 0.0 + 1. Cm

The difference between the two equations for the resistors considered here is no more than 0.01 ppm.

In addition, a least squares fit of a straight line was made to the differences

of duplicate measurements from both sys- tems. Ideally one would expect no cor- relation between the differences; i.e., the

slope of the line should be close to zero. The least squares fit yielded the following

line. Da = 0.03 — 0.31 Dm

where Da = difference in ppm of dupli- cates from the automated

system

Dm = difference in ppm of dupli- cates from the manual

system

The residual standard deviation was

0.021 ppm and the standard deviation of the slope was 0.277 ppm.

In addition, a fit was made of a zero

slope line and the following resulted. Da = 0.024 The residual standard deviation was

0.021 ppm.

These results indicate that there was no

correlation between the differences from

both systems and that differences of the automated system are larger than those of manual systems by 0.02 ppm at the

one sigma level.

LCS 700:70:991 ASQC 11/76-245 References

(1) Wells. T.E., and Gard, E.F., "An Inte- grated System for the Precise Calculation of Four-Terminal Standard Resistors." IEEE Trans, on Istr. and Meas.,

Vol. 1M-20, No. 4, (Nov. 1971).

(2) Natl. Bur. Stand (U.S.), Tech. Note 990, 42 pages (Aug. 1973).

Acknowledgements

Construction of the crossbar switch by R.E. Long, programing and data reduction by D.E. Bruch, and R.C. Fronk, assistance in making measurements by L.R. Cooke, and

valuable suggestions and criticism by N.C.

Belecki and T.E. Wells are gratefully

acknowledged.

349
Reprinted by permission from IEEE TRANSACTION'S ON

INSTRUMENTATION AND MEASUREMENT

Vol. IM-20. No. 4, November 1071

Copyright © 1971, by the Institute of Electrical and ETecTrmiics Engineers, Inc. PRINTED IN THE U.S.A. An Integrated System for the Precise Calibration

of Four-Terminal Standard Resistors

THOMAS E. WELLS and EARL F. CARD, member, ieee

Abstract—A description is given of the system currently in use

at the National Bureau of Standards for measurements of 1-Q standard resistors of the Thomas type. A tenfold improvement in accuracy over the former method has been realized. Resistors of this type are now reported to eight decimal places with a total uncertainty of 0.08 ppm. The latter figure includes a three-standarddeviation limit for random errors plus an estimate of systematic

errors [1]. The new system is not only more rapid in operation than the old one (fewer man hours required), but has consistently produced data

of superior resolution while operating with a lower test power in resistors. A dc current-linkage balance is used to determine the standard-unknown differences while both are connected in series and totally immersed in a specially designed circular oil bath, which remains completely covered during all tests. Bath temperature is maintained constantly at 25 000°C (±0.003°C) in a unique manner.

Manuscript received May 27. 1971; revised July 9, 1971. The authors are with the Electricity Division, Nation.nl Bureau

of Standards, Washington, D. C.

Introduction

IN 1963, as reported to the clients of the National

Bureau of Standards, the precision of the resistance

of double-walled resistors was increased from 1 to 0.1 part/million (ppm). Since this precision was near

the limit obtainable on the bridge in use at that time

(the Wenner bridge [2]), it seemed prudent to plan a new resistance measurement system capable of meeting

expected demands for increased precision. It was planned

that this new system would not only be capable of greater

precision but, if possible, would be faster and easier to

use.In developing a new system, the following recognized

limitations of the old system were kept in mind: 1) the requirement for manually made lead adjustment in the

ratio-arm connections; 2) disturbance of the temperature

350
IKKE TKINS.MTIONS US I N STKU.M KM AT10N AND MKASI'HKMKNT, KUYKMIIKK 1971

equilibrium by handling of the resistors when performing

interchange; 3) generation of transient thermal EMFs

when changing potential connections to the resistors; 4) uncompensated errors in the ratio set itself caused by

unequal power coefficients of certain ratio resistore; 5) temperature-control difficulties partially inherent in the

original bath design such as its rectangular shape and

placement of heater lamps close to the measured resistors. Lack of a cooling element rendered the bridge

inoperative when the ambient room temperature rose above 24.0°C. Temperature control was ±0.01 °C at

best, but frequently averaged ±0.02°C; and G) electrostatic disturbance caused by charged objects, e.g., people

moving around the room and electromagnetic disturbance caused primarily by the heater lights in the bath turning on and ofT. Although most of these limitations of the old Wenncr

bridge system could have been eliminated by the construction of a modified bridge system, it was decided to use a potentiometric system. This method seemed at- tractive due to the development of the direct current comparator by Kusters and others at the National

Research Council of Canada [3], [4].

Integrated System

Constant Temperature Bath

A tank was constructed large enough to accommodate

a circular mercury stand holding 15 Thomas-type re- sistors in a series ring. Salient features of this bath are the following.

1) It is cylindrical in shape with approximately 550-1 capacity. The radial oil flow is produced by a belt-driven

propellor located in the bottom center of the tank. The oil flow moves horizontally between two cylindrical screens, a central one (15 cm diameter) of fine mesh capped at the top and an outer very coarse mesh screen (Fig. 1). The

oil return path to the propellor is via a false bottom

containing coiled tubing designed to form a heat ex- changer. The oil circulation is superior to that obtained

in the rectangular tank of the Wenner bridge.

2) The resistor ring stand is positioned so as to place

all resistors at the same middepth oil level and also equidistant from the central oil distribution screen.

3) No electrical heaters or coolers are installed in the

550-1 bath. Temperature control is obtained by regulating

the circulation of cooling oil pumped from a smaller

(110-1) auxiliary bath through the heat-exchanger tubing

located in the bottom of the larger tank. This cooling

oil carries away the heat produced by the stirring of the

oil in the large bath. It was found that variations of 0.02°-0.03°C in the temperature of the auxiliary bath

would be reflected by variations 1/10 or less as large in the

larger bath. The temperature of the bath can be controlled manually

by varying the flow of cooling oil by adjusting a needle

valve in the oil pipeline. Alternatively, the needle valve can be set high enough to assure cooling of the bath and

the flow of oil can then periodically be interrupted by

a valve in the cooling oil circulating line that is controlled by an electronic sensor in the large bath.

Current Comparator

The direct current comparator operates on the principle that when two magnetomotive forces imposed on a magnetic core by currents in two windings are equal and

opposite, the flux in the core is zero. This balance condition can be detect ed by a modulation detector. The

detector circuit can be used to control the output of a

current source, connected into one of the windings so as

to automatically maintain the condition of zero flux in the core. How the current comparator can be used to compare resistors can be seen from the equation developed

below, with reference to Fig. 2. The basic equations are as follows when

Nv , N, primary and secondary number of turns;

/„, /, primary and secondary currents; Rn, Ii,, R. resistances of "dummy," unknown and

standard.

At flux balance in the transformer core, NJ, = NJP

or

/.//„ = NJN. (1)

At voltage balance the IR "drop" across the "dummy"

(secondary) resistor R„ and the IR drop across the

selected unknown resistor Rx in the primary circuit results in RJV = I„R„

where /„ is the secondary current at balance or

R, = (I.,/I„)R„-

Substituting turns for current, from (1) we have

Rx = (NP ,/N.)RD (2)

where TV,, is the number of turns in the primary winding

at balance.

Similarly when a resistor of known resistance R, is being measured we have

R. = (NV ./N.)R„. (3)

Dividing (2) by (3) we then have

R. = {NP ,/N,.)R.. (4) Measuremenl Teclmique

Briefly, the technique is one in which the voltage drop

across each of the client's resistors is compared indirectly

(via a stable dummy resistor) with the voltage drop

across each resistor in a reference group while all arc connected in series and totally immersed in the oil bath 351
WELLS AND CARD I CALIBRATION OF FOUR-TERMINAL RESISTORS now

CONTROL

ASY.SAY-. —II rr

I

I

.

|

| LEVEL |_ __ -— /T

\,—5—,1

I

i

'////////.'//

' /;'///>/,; ;','/,

cxyoi'.-ozvo::

CIRCULAR BATH

THERMALLY INSULATED 550 LITER

AUXILIARY BATH

IIO LITER

LfVO.

H H

COOLANT ^

Fig. 1. Two-stage temperature control system.

maintained at 25.000°C ± 0.003°C. The dc-current comparator (ampere-turn balance) is used for making the comparisons while all the standard and unknown re- sistors are carrying the continuous test current of 0.1 A

(0.01 W/Thomas resistor). The comparator is usually

used near the maximum setting of the balance dials (2000

turns of the primary winding) and the circuit "trimmed"

in such a manner that only the last three or four dials are changed in balancing. Final balance settings of the seven decade controls are recorded and utilized solely to obtain accurate standard-unknown differences. A series of balances between the 0.5-Q dummy and

each of the standard and unknown resistors in turn is made by adjusting the seven dials controlling the turns

of the primary winding. Sensitivity is sufficient to readily discern 0.5 step of the seventh dial. As the first dial has

20 steps and all balances in this instance are made with

it at the twentieth step, the 0.5 step is 0.025 ppm.

With the 0.1-A dc primary current automatically controlled and the slave supply tracking it at 0.2-A dc (the zero flux feedback ensures this condition), the series of balances between the 0.5-0 dummy and of the standard

and unknown resistors in turn can be made as rapidly

as the technician can operate the potential selector switch

and adjust the seven dials controlling the taps to the primary winding using the standard technique of reversing the currents to eliminate the effect of "parasitic" EMF

in the d&ectcr circuit. Standard operating procedure at NBS is as follows.

1) The bath is "loaded" with a maximum of eight

clients' resistors on the day preceding measurements,

oil depth checked, cover replaced, and 0.1-A current 52
1KKK TI<\\S\rnONS ON INSTHI Ml STATION \M> Mi;\SI III. Ml ST. MIVI MHI.II I'.lTl

FEED BACK.

OIL LEVEL

D.C. AMPERE-TURN

BALANCE

OIL LE.VEI

Fig. 2. Circuit for comparison of resistors.

switched on. Note that the bath is in constant temperature control with continuous stirring even when not

in use.

2) On the following day, the bath temperature is checked, trimmed slightly if necessary, and the temperature recorder started for the calibration run.

3) Balances are then made as described for each of the

15 selector switch positions, 1-15, and dial readings

recorded.

4) Room barometer is read and pressure recorded along

with room temperature.

5) Repeat balances are then made in reverse order, 15-1.

6) Approximately five hours later, without opening the

bath, steps 2-5 are repeated entirely by another technician.

7) Test current is turned off; bath is opened and

unloaded. We now have four balances for each switch position.

Averaging these and knowing the average correction of the five NBS reference resistors, only simple arithmetic

is necessary to arrive at the proper corrections for each

of the unknown resistors. The value of the dummy

0.5-12 resistor is unimportant as it does not enter into the

calculations. However, its resistance must remain constant during a given run. A low power coefficient is desirable for the 0.5-12 resistor although in this case the secondary current is essentially constant because (inly small changes are made in the primary turns setting. In addition to the five reference standard resistors, there

are always present in the series an extra pair of NBS

Thomas 1-12 resistors that are measured and serve as controls in the data analysis. Resistors of the double-walled type such as these are

affected slightly by pressure variations [5], [6]. Therefore,

if maximum use is to be made of the reported values, the

effect of pressure must be considered. Although the temperature and oil-immersion depth arc held constant, the ambient air pressure will naturally vary; therefore, before a mean value for the correction of the five can be utilized,

it must be adjusted back to tin' 101 325 K m' pressure

353
IKI'l: TKVNSW I'lONS ON INSTUI M KNT \ THIS (Ml MKA-SUtKMIvST, VOL. IM-20, Ml. 4, NOVKMBKIt 1071

base, from (he total pressure as measured in a given

calibration run.' Pressure coeflicient tests were carefully

made on each of the reference resistors at NBS. Results have shown the effect to be positive and linear with no

appreciable hysteresis.

Clients' resistors whose individual pressure coefficients are not known in most cases are simply reported at the

total pressure existing during the testing period. An

approximate figure for commercial units is 0.000 015-ppm

decrease in resistance per newton per square meter de- crease in pressure on resistor (multiply pressure in milli- meters of mercury by 133.32 to obtain pressure in newtons

per square meter). For a user located at an altitude of one

mile the resistance of this type resistor will be about

0.3 ppm less than the value at the altitude of the NBS

Gaithersburg Laboratory.

System Performance

The resistance values of Thomas-type standard resistors submitted to NBS for calibration prior to July 1, 1970, were reported to 0.1 ppm with an uncertainty of ±1.0

ppm. Since that date, using the new system the values have been stated to 0.01 ppm with an uncertainty of only 0.0S ppm. The latter figure mcludes a three-standarddeviation value for random error due to minute temperature, pressure, resistance fluctuations, and lack of resolution plus estimated amounts for the uncertainty in temperature and pressure measurements, the only known

sources of systematic errors. The test power, formerly 0.05 W/resistor (intermittent)

1 Total pressure existing at tlie level of the surfaces of the amalgamated current terminals when the resistor is immersed so

as to place the terminal surfaces 60 mm below the oil level.

is now 0.01 \V (continuous). The performance of the oil batli now permits specification of temperature at which

the calibrations are performed as 2o.00° rather than

2o.0°C, as was previously the case. The new reports

state the total pressure on the resistor during calibration. Additional advantages of the new system include: elim- ination of the tedious and time consuming lead balances that were necessary with the Kelvin double-ratio circuitry;

and elimination of a secondary or working group of NBS

standards. Clients' resistors are now compared directly

with the National Reference Group of standards. This measurement system was completed in 1969 and

has been in successful operation since that time. The

initial circuitry was designed solely for comparisons

of Thomas-type standard resistors. Experiments indicate that with proper modifications the same type of system

can be extended to perform one to one comparisons of

resistors in the range 0.1-100 ft and such extension is contemplated.

References

[1] N-at. Bur. Stand. "Improved accuracy in Thomas-type one- ohm standard resistor calibration," Tech. Ncux Bull., vol. 54. 1970, p. 188. [2] F. Wenner, "Apparatus and procedures for the comparison of

precision standard resistors," J. Res. Nat. Bur. Stand., vol.

25, 1940, p. 229; also reprinted in NBS Handbook, vol. 77, 1961. p. 1. [3] M. P. MacMartin and N. L. Kusters. "A direct-current-com- parator ratio bridge for four-terminal resistance measurements." IEEE Trans, lustrum. Men*., vol. IM-15, Dec. 1966. pp. 212-220.

[4] X. L. Kusters and M. P. MacMartin, "The application of the

direct current comparator to a seven-decade potentiometer."

IEEE Trans. Instrum. Meas., vol. IM-17, Dec. 1968, pp. 263-

268.

[5] J. L. Thomas. '"Stability of double-walled Manganin re- sistors." J. Res. Nat. Bur. Stand., vol. 36. 1946, p. 107. [6] Nat. Bur. Stand., "General information of standard resistor?. (Elec. Ref. Stand. Sec. 211.01), Dec. 1970.

354
Reprinted by permission from IEEE TRANSACTIONS ON

INSTRUMENTATION AND MEASUREMENT

Vol. IM-19, No. 4, November 1970

Copyright © 1970, by the Institute of Electrical and Electronics Engineers, Inc. PRINTED IN THE U.S.A. NRC 11548 Resistive Voltage-Ratio Standard and Measuring Circuit

RONALD F. DZIUBA and BERNADINE L. DUNFEE, senior member, ieee

Abstract—This paper describes a highly stable, guarded dc

voltage-ratio standard and the measuring network and techniques used to establish the values of its ratios to an accuracy of 0.2 ppm. The entire system is housed within a dry-air enclosure whose temperature is maintained at 23 ± 0.05 °C. Discrete ratios from 1:1 to 1000:1 are provided, with maximum rated voltage set at 1000 volts. The design of the standard was chosen so that a redundancy of measurement could be incorporated in the system. Thus each successive

ratio is measured by a substitution or "bootstrap" method and by

satisfying the conditions of the series-parallel principle, the 10:1, 100:1, and 1000:1 ratios are measured by a second independent method. The design also admits additional checks on the validity

of the measurements. An analysis of measurement errors and a discussion of their possible origin are included. Since the intent

also was to design the ratio standard for low-frequency operation some preliminary data are included on its ac performance.

Introduction

RECENT years have brought challenges to the

scientific and technological communities that were barely discernible a few decades ago. Advances made in meeting these have been possible only

through the more precise and accurate measurement

of physical quantities that often are interweaved within

vast complex measuring systems and often cover a range

Manuscript received June 4, 1970. The authors are with the Electricity Division, National Bureau

(if Standards, Washington, 1). C.

of magnitudes of several decades. Measurement of dc

voltage from the 1-volt level of the standard cell to 1 or 2 kV is a case in point—a necessary and important

link in the system. Extension from the approximate

1-volt level is realized through ratio networks that most

generally are built up from resistive elements. Those

that can be more aptly classed as ratio standards contain

a limited but adequate number of discrete ratios and

are most often used in comparison circuits to assign values to working standards of somewhat lower accuracy

class. The ratio standard that for many years performed

this function at NBS was a Silsbee design [1]. It featured

a separate guarding network for the more critical sections, exhibited only small changes with self-heating, and

admitted to a self- or step-up calibration process. The

stability and the accuracy to which its ratios were known

were of the order of 5 ppm of ratio. Some degradation

in accuracy followed when using it in a comparison

network to assign values to other types of ratio networks.

The need for an improved standard, with a more

accurate assignment to its ratios has become evident. Voltage supplies of constant known output and digital voltmeters for monitoring voltage are approaching

claimed accuracies of 10 or 20 ppm. These must be

evaluated through ratio networks, leading ultimately

355
DZH'BA AND DUNFEE: RESISTIVE VOLTAGE-RATIO STANDARD

to an assignment in terms of the national standard of EMF. Thus, ratios must be known with increasing accuracy at each higher echelon of measurement thereby

placing stringent requirements on those ratio measure- ments at the national level. To help provide the increased accuracies required in such a chain of measurements, the intent at NBS was

to design a more versatile standard of improved stability

and with ratios more closely equal to their nominal

values, together with a measuring network and technique

for assigning those values of ratios to an accuracy better than 0.5 ppm of the output (5 X 10

-10 of the 1000-volt

input).

During this study and prior to the final design described

in this paper, several ratio networks were considered. One, which looked promising, was developed and ex- amined as an experimental model DD-2 before it was

rejected for practical reasons. Theoretically, the design

admits errors only of second order but the problems

encountered in its construction, operation, and in the assignment of its errors outweighed the advantage

predicted by theory. However, the basic design, although not new, might be useful in some applications and provide

high accuracy where a model of limited range or one

with few ratios is desired. For this reason, a brief dis- cussion is included in Appendix I. Rejection of this design led to the one described that

is simpler in construction, more versatile, and includes a redundancy of measurement feature. Study of an

experimental model (DD-3) built in 1967 proved its feasibility and furnished the guidelines for its design

for both dc and low-frequency ac operation. Further

advantage was gained by providing the ratio standard

with its own measuring network so that the combination

would serve as a measurement system. This paper

describes the measurement system, identified as model

DD-4, and includes an analysis of the measured ratios that led to an estimate of 0.2 ppm for the uncertainty.

It is interesting to note that a similar basic design

for a ratio standard was being explored simultaneously

and independently by Ohlon [2].

Basic Circuits

The measurement system as regarded in this application

includes two networks housed within the same enclosure. One is the ratio standard, the second is the main bridge array used to measure its ratios. Each is electrically

separate from the other and either is available for use with external circuitry. The voltage supply, detector,

and connecting leads are the only external accessories required. The system is shown schematically in Fig. 1 and its physical arrangement appears in Fig. 2.

Ratio Standard

The "working" branch of the ratio standard designed

for 1000 fl/V is indicated in the schematic circuit of

Fig. 3. Carefully selected resistance elements are arranged

singly or in combination to form 28 sections. Taking

the first section as the reference with a nominal value

of R = 1 kS2, the remaining sections divide into three groups A, B, and C, each containing nine sections. Those

of group A have a nominal value of 1 kS2 while those of groups B and C have nominal values of 10 k£2 and 100

kfl, respectively. In the series mode as shown, 27 ratios (referred to the first section) are available, extending over three decades to a maximum of 1000:1. The resistance of each section is nominally equal to either the resistance

of the preceding section or to the summation of all preceding sections. As stated in [1], this satisfies the

criterion by which such a network is judged to be self- calibrating. Thus, the bootstrap technique described in

[1] is applicable and requires only the measurement and

appropriate summations of small differences between

nearly equal quantities. In addition to the series mode, the nine sections comprising a given group are geometrically arranged for easy transfer to a parallel configuration [3]. Links, or

their equivalent (indicated by dotted lines), from o to b and from c to d reduce group A to a parallel resistance

of 1 kS2, nominal; similarly, those bridging groups B

and C reduce these groups to nominal values of 10 and

100 kfi, respectively. Inclusion of the paralleling feature

increases the number of available ratios with some

convenient duplications, but more important, it makes

possible an intended redundancy of measurement so that more accurately known ratios are assured. The ratio standard also carries a "guard" chain of 200 fi/V arranged and mode operated in similar fashion

to that of the working branch. Operating in parallel

with the working branch, its function is twofold: 1) it maintains the proper potentials on metal shields that surround the critical parts of the working branch and

prevents the loss of current through the insulation,

2) it completes the shielding network of the measuring

circuit and, through its accessible terminals, maintains

the branches of this network at the proper potentials, so that leakage paths through insulation are diverted from the measuring circuit.

Measuring Circuit The measuring network, although accessible for external

use, is designed as an integral part of the system for the ready calibration of the ratio standard. With suitable connecting leads, it accomodates the ratio standard in either its parallel or series mode and can be used entirely or partially depending on the calibration method employed. The circuit as it would be used for either mode

of the standard is indicated schematically in Fig. 4. Ratio arms A and B operate on the principle of the

direct-reading ratio set (DRRS) [4]. Resistor B and

resistor A, when at midscale setting, have nominal

values of 1 kft, with A adjustable by 50 ppm on either

side to a least count of 0.1 ppm. (Interpolation extends

the resolution to 0.01 ppm.) Three resistance elements

having nominal values of 1, 10, and 100 kft are separately

356
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, NOVEMBER 1970

rl B .2k A

r3 -201(0

Fig. 1. Model DD-4 voltage ratio standard and measuring circuit.

Fig. 2. Physical arrangement of DD-4. A—Plexiglass top pieces. B—Card-type resistor. C—Coaxial terminal. D—Brass guarding

plates. E—Guard resistors. F—Folded Cu 4-terminal junction. G—Brass guarding rods.

available as S. The R arm is occupied in sequence by

the sections of the standard whose differences are being

determined. Since the bridge measures only small dif- ferences between nearly equal quantities, only moderate

accuracy of measurement is required.

The guarding network is essentially a mirror image

of the main bridge. With the shields maintained at proper potentials, leakage currents are directed from

shield to shield and confined to the guarding network.

Design Parameters

Ratio Standard

The performance of this type of ratio standard is governed in great measure by the behavior and structure

of the resistance elements that comprise the working

branch. These were carefully selected to reduce the

largest sources x>f error to manageable size. Special effort was made to acquire elements with the

(a) (b)

Fio. 4 (a) Simplified measuring circuit for a single section

(b) for a series-parallel array.

lowest possible temperature coefficient. Measurements

and proper selection assured a temperature coefficient

less than 0.5 ppm/°C for the 1 and 10 kft elements. Thus, each section of groups A and B (Fig. 3) comprises

a single element. A similar selection provided 100-kQ

elements with temperature coefficients not exceeding

1 ppm/°C. Allotting four elements to each section of group C and arranging them in a series-parallel network

with some matching of positive and negative coefficients reduced the temperature coefficient per section to less than 0.5 ppm/°C.

A related and equally troublesome source of instability

and uncertainty arises from temperature changes produced

357
DZIl'BA AND DUN FEE : RESISTIVE VOLTAGE-RATIO STANDARD

in the elements when under load. This can be alleviated by designing for the lowest power consumption except

that an upper limit is imposed on the value of resistance that can be chosen. Leakage currents and their control become paramount problems so that some compromise

is required. The design for 1000 ft/V offers a low operating

current without placing too stringent demands on the guarding system, the sensitivity of detection, and the

stability of the resistors. Further reduction of uncertainties associated with heat flow was assured by selecting

single-layer card-type elements of Evanohm with each

having a radiating area of 16 cm

2 per surface. Each

element is suspended between its respective terminals with its two surfaces exposed to the coolant and removed

sufficiently from neighboring elements to alleviate problems from proximity heating.

The nine elements comprising a given group were

closely matched (adjusted if necessary) to satisfy the requirement that the proportional correction to the

resistance of a group when in the parallel mode be identical to that when in the series mode within the accuracy

sought.

The high accuracy and stability prescribed for this system dictated that complete guarding along the working

resistance chain be provided. The departure of each

guard section from its nominal value of resistance does not exceed 0.01 percent so that the guarding sleeves and the working terminals they enclose are essentially at the same potential. Thus, with guarding and high

insulation resistance, errors caused by leakage currents

are less by several orders than the accuracy prescribed

for the measurement process. To prevent heat generated

in the guard from influencing the working circuit, the guard chain is mounted in an upper channel external

to the enclosure and open to the outside. Measuring Circuit The DRRS, although available to external circuits, was essentially tailored to the application at hand. The

adjustable arm is a small and compact commercial

unit with Waidner-Wolff circuitry and card-type resis- tance elements. The resistor that forms the B arm and

those that occupy the S arm (Fig. 1) in sequence are

also the card type having been selected by the same

criteria and mounted in the same fashion as those of the ratio standard. Additional Considerations

Terminal Junctions

Proper selection and arrangement of elements that

satisfy design criteria neither guarantee that the values

assigned to the ratios are conclusive nor that they are

valid when the standard is used to calibrate other ratio networks. Unless other precautions are taken, values obtained by two or more independent methods of equal

precision could differ by significant amounts with no

assurance, which, if any, represents the value of the

ratios as used in calibration. Thus, special attention

was given to the design of the terminal junctions and

provision was made so that effects from their inter- connections and their connections to both internal and

external measuring circuits could be predicted and

verified experimentally.

An ideal junction is four terminals, designed and adjusted

so that all four-terminal resistances are zero [5]. This

is the preferred junction, since, if four-terminal techniques

are used to measure a chain of nominally equal resistance elements, it defines each resistance precisely. Further- more, through the use of compensating fans or their equivalent, it permits the parallel arrangement of these elements without introducing spurious resistances through

the paralleling process. This subject has been covered

quite thoroughly in the literature [6], particularly the use and analysis of compensating fans as proposed by

Page [7]. If the ideal junction and compensating fans were applied to the circuitry discussed in this paper as indicated in Fig. 5, no discrepancies would exist between

the identity of the measured ratios and those in use. However, a study of the preliminary experimental

model (DD-3) indicated that strict adherence to the geometry of the ideal junction (three terminals symmetrical to a fourth) was not necessary even though the measuring technique selected would not be strictly four terminal. For these and practical reasons, each

junction is a small thin copper plate with its terminals

located at the four corners of a square (see Fig. 2). The

plug-type coaxial terminals are silver-plated brass with

polytetrafluoroethylene (PTFE) insulation. The plug- socket resistance of a connection is approximately 450

and the contact resistance is repeatable to within

±10 yufi. The four-terminal resistances of each junction measured

5.5 ± 0.5 n$l for both the direct and cross resistances. The maximum uncertainty from junction effects and

from possible variations in lead and connector resistances was estimated at less than 0.05 ppm. Appendix II indicates how this estimate was approached. The results and

their analysis in Appendix III indicate the experimental

verification.

Detection System

The detection system consists of a photoelectric gal- vanometer amplifier (PGA) whose unbalance signal is further amplified and displayed by an electronic detector. To provide proper negative feedback for the PGA, a 500- ohm resistor is connected across its output terminals. The

system is operated at a sensitivity of 30 nV/mm. The

noise level is less than 40 nV peak-to-peak during the time period for a single measurement. By visually inte- grating the signal voltage over a time period of approximately 3 seconds, it is possible to detect 0.01-ppm changes

in the bridge circuit at the 1-mA level. The metal cases of the amplifier and electronic detector

as well as the shields of all leads in the detection system

are driven at the appropriate guard potential. Thus,

the integrity of the insulation is maintained and effects

358
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, NOVEMBER 1970

Fig. 5. The parallel mode of the ratio standard, emphasizing

ideal junctions and their interconnections. The unlabeled resist- ance elements are lead and junction resistances.

caused by leakage are reduced to a negligible level. In addition, electrostatic interference is suppressed. The

detection system is isolated from the 60-Hz power circuits to reduce noise problems.

During a measurement operation, the DRRS is adjusted

while the bridge current is periodically reversed until a null condition results.

Controlled Enclosure The ratio standard and the measuring network are immersed in a dry-air bath maintained at a temperature

of 23 ± 0.05°C. Heat transfer occurs by forced convection

using two fans that provide laminar flow across the

resistance elements. The thermally lagged bath of ap- proximately 0.1 meter3 consists of two aluminum boxes

separated by 2.5 cm of foam polystyrene and two plexiglass top pieces separated by a 2.5-cm air space. The temperature control system consists of 1) a thermoelectric heat-pump assembly that extracts a constant amount of heat and 2) a proportional control circuit that regulates the power input of two (0-15 watts)

heater elements. Twelve copper-constantan thermocouples strategically located within the bath monitor

the temperature to within ±0.05°C.

A recently installed regenerative drying system holds

the humidity constant at 4 percent relative humidity.

A small amount of filtered and desiccant-dried air under

pressure is continually forced into the bath while two

small ports located at the opposite end of the bath serve

as the exhaust. Methods of Measurement

Four essentially independent methods were used during

the period of evaluation and several lead assemblies were introduced for each method. Two of these were

considered as the principal methods while the others

served as occasional checks. The one identified as the bootstrap method is regarded as primary since values

are obtained for each consecutive ratio. The other identified as series-parallel serves as the redundant method,

providing additional values for the 10:1, 100:1, and

1000:1 ratios. Although both techniques utilize the

built-in measuring circuit they are sufficiently different to be regarded as independent.

Bootstrap

The circuit of Fig. 4(a) shows the exact arrangement

of leads and shielding as used in the bootstrap method

with one section of the standard (e.g., the reference

section), in the R arm. The l-k£2 sections, beginning

with the first or reference section, are introduced in the R

arm in rapid succession and the bridge is balanced for a

null on the detector. The balance equation for each

k section is of the form

K1>4 = + D1 . t - D0 ] (1)

where Si has the same nominal value as Ri, h . The reading on the DRRS stated in proportional parts is Z) ltt while

D0 corresponds to the reading that would exist for an

exact 1:1 ratio of A:B. (The derivation of (1) appears

in [1].) The correction to each successive ratio through

10:1 is obtained by taking differences between each

equation and the reference equation and averaging their successive sums. The summation X**=i° ^i.* *s the resis- tance of the first decade that is nominally equal to the

resistance of each section in group B (Fig. 3). A comparable set of 10 balances is then made beginning with

the first decade and continuing through the B group

with Si changed to S2 = 10 kS2. A third set of similar balances carries the measurements through the 1000:1

ratio. Under rated conditions each measurement is made

with 1 mA through each section. The general equation

from which any ratio (referred to the first section) can

be computed is Sm .„ Rui

(2) where Sm , t is the sum of all resistances up to the kth

tap of decade m, and RlA is the reference section. Each

term in the parenthesis is the sum of all the measured

differences throughout a decade while the last term

accounts for the measured differences up to the k section

of interest. (The derivation of (2) is similar to the one

appearing in [1].) Since /cl0m_1is the nominal ratio,

(2) is of the form

N = Nn[l + c)

where c is the correction to a ratio in proportional parts. 359
DZIUBA AND DUNFEE I RESISTIVE VOLTAGE-RATIO STANDARD

It is important to emphasize that a systematic error that might exist in each measurement is not cumulative

within a decade since it, like the differences, partakes

of the averaging process. Any accumulation occurs only when passing from one decade to another.

Series-Parallel This method provides corrections to ratios 10:1,

100:1, and 1000:1 and requires six measurements in contrast to the 30 of the previous method. In each successive set of two measurements, each group (A, B,

and (?) is converted, in turn, into its parallel mode and

compared against all the series-connected sections pre- ceding it. Thus in the first set of two balances, the ref- erence section occupies the R arm and is followed by

group A connected in its parallel mode, as shown in

Fig. 4(b). In the second set, with group A returned to

its series connection, the first decade followed by group B in its parallel mode occupies the R arm. A similar sequence repeated once again carries the measurements

through the 1000:1 ratio. For each successive set, S

takes on the values 1, 10, and 100 kU. Three sets of

differences and the average of their successive sums

with the series-parallel principle applied yield values

for the three ratios. The general equation for any ratio (referred to the first section) is Nm = 10" 1 +

where m is the number of the decade and

p-1

is the sum of the measured differences. Two principal lead assemblies were used. In the one

case the paralleling of groups was effected by using both

current and potential fans. In the other only one pair

of compensating fans was used. No difference in the

results was noted so that the simpler arrangement was

preferred. This is further indicated in Fig. 9 of Appendix

II where the connections for both the reference section and the parallel array are shown.

Additional Methods

The two "check" methods alluded to earlier are identified as the "interchange" and "Kelvin bridge" methods.

In the former, only arms A and B of the internal bridge

are used. The other two arms of the bridge are occupied

in turn by two adjacent sections with each pair advancing

by one section. Since decade (m) forms the first section

of decade (m + 1) the process can be carried forward

through the remaining sections. In this kind of measurement, D0 must be determined by the usual interchange

technique. The measuring process is similar to that of the bootstrap method in that a value is obtained for each ratio. The second method, as the name implies, is a four- terminal measurement. The step-by-step procedure of

the bootstrap method is followed but the sections art measured as four-terminal resistances using a Kelvin

bridge. This method was used only for the 1-kft decade

where effects from terminal junctions and lead arrange- ments would be greatest.

Results

An analysis of all dc data accumulated from December

1968 to November 1969 is contained in Appendix III. Data taken since the completion of the analysis and too

late for inclusion continue to substantiate the conclusions

as stated. The dc data obtained by the two principal methods

covered 50-60 complete sets of measurements for each

of two applied voltages. (Additional complete or partial

sets obtained on occasion from the two supplementary

methods totalled 14). The initial study covering about

six months was made at 50 percent rated voltage. When

evidence indicated that the measuring system was suffi- ciently stable and that measurements probably could be made to better than the 0.5 ppm anticipated, the system was moved to a location where ambient influences could be better controlled. The concentrated study

continued for 4 months with measurements made at the reduced voltage over a 2-month period followed by those

at rated voltage.

Analysis of the data at rated voltage shows that for a 99 percent confidence interval no difference greater

than 0.1 ppm exists between the two methods. This

excellent agreement between two independent methods

makes it very unlikely that a systematic error greater

than 0.1 ppm has gone undetected. Additional information is available from Fig. 11 where

the results at rated voltage are plotted against elapsed

time for the 10:1, 100:1, and 1000:1 ratios. Fig. 11(c)

is interesting since it appears to indicate a difference approaching 0.3 ppm between the two methods. However,

as shown in the appendix, about 0.2 ppm arises because

the power level at which each section operated was

appreciably different for the two methods. Hence, this portion of the observed difference simply reflects a dif- ference in test conditions and can not be regarded as a systematic error that lies hidden in one method or the

other. Furthermore, the load conditions on the sections at rated voltage for the bootstrap method are essentially the same as those that would exist when in use, so that

this difference could be regarded as a systematic error only if the series-parallel method were used alone to assign values at so-called rated voltage. Adjustment

of the values to account for the known change with

load led to the assignment of 0.1 ppm as the systematic

error through the 1000:1 ratio. As justified in Appendix III, greater weight was

accorded the values obtained at rated voltage. The

stability of the system improved after its transfer to the inner room and the subsequent analysis indicated that the standard deviations at 50 percent rated voltage were reduced, the change being appreciable on the 1000:

360
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, NOVEMBER 1970

ratio, and were more consistent with those at rated

voltage. Also toward the end of the measurement series at reduced voltage, when the values were scrutinized more closely, it became evident that humidity effects, though small, could no longer be considered negligible.

These are discussed more fully in the appendix; however,

they are evident in Fig. 11 where the changes in the drying agent are noted along the abscissa. Evidence

indicates that much of the random behavior arises from

this cause. Data accumulated after an improved humidity

control was installed have produced no evidence to the

contrary. There are two known mechanisms that could produce the humidity effects observed: 1) surface leakage

across the resistance elements and 2) forces on the fine wire caused by dimensional instability of the wire insula- tion or mica former. Leakage across the insulation between the guard and working circuit appears negligible.

An estimate of a for the first decade where effects from

humidity are least is 0.05 ppm. This includes not only

the random error of the measurement process but also any random changes in the ratio standard over a period

of about 2 months. Taking this value as applying also to the other decades (under dry conditions), an uncertainty

is assigned at 0.2 ppm, based on a Za confidence limit

for random errors with a 0.1-ppm estimate for the residual systematic error. AC Characteristics

Several features were incorporated in the design of the ratio standard for operation at 100 Hz or below. These include:

1) the use of single-layer card-type resistors having

small residual inductances and shunt capacitances;

2) the use of air as the heat-exchange fluid for the bath instead of oil or a similar fluid having a larger

dielectric constant;

3) the design of junctions that could be guarded easily,

along with the use of guarded connectors as the terminations of these junctions;

4) a low guard resistance (200 Q/V) to minimize the

effect of external shunt capacitances on the guard ratio and thus further reduce their effect on the main ratio [8]; 5) the use of brass rods, connected to appropriate

guard-circuit taps, at various points to suppress errors caused by capacitive currents from the resistors to their surroundings.

The ratio standard was calibrated at 100 Hz by the

substitution method using an inductive voltage divider. The output of an auxiliary inductive voltage divider connected to the common point of the guard-circuit

resistors in the bridge provided for the phase adjustment.

The calibration procedure is similar to that described

previously in the bootstrap method. Results of these

tests compared to the dc values agree to 0.1 ppm for

ratios 1:1-10:1, 0.2 ppm for ratios 10:1-100:1, and 2.5 ppm for ratios 100:1-1000:1. The good agreement of the lower ratios further substantiates the accuracy of the dc values. The larger disagreement of the 100-kfl

3

sections is due probably to the uncompensated residual shunt capacitances and to significant dielectric absorption

of the mica winding form of the resistors. The above ac data are only preliminary and further study is planned.

Summary

A stable voltage-ratio standard with extremely small

errors and the networks for measuring its dc ratios to an accuracy of 0.2 ppm were described. Preliminary

study of the standard at 100 Hz indicates a comparable

performance can be expected at low ac frequencies.

Investigation of the ac performance will continue. This

development can lead to a more accurate assignment

of ratios and relatively large dc voltages throughout

the entire chain of measurements.

Appendix I In the experimental model DD-2, a "base" element,

which provides a reference ratio, contains two sets of n

nominally equal resistors connected in series [9]. Each

set is arranged so that its n resistors can be connected

either in series or parallel. Two ratios are possible. A

ratio Ni 1 + n

2 (l + a, — /3P) is obtained by connecting

the a set in its series mode and the /3 set in its parallel

mode. A second ratio N2 1 + n

2 (l + /S, — ap) is formed when each set is reconnected for its alternate mode and the two are interchanged in position. (The a and 0 terms are small corrections to the resistance of the respective sets as they occupy the series or parallel

position.) If the series-parallel principle is satisfied a. = av and j8p = /S, and the mean ratio N 1 + n

2

. Thus a reference ratio is provided that, theoretically

and to first order, is a function only of the number of elements. Extension of the principle to higher ratios

is effected by adding sections whose resistances could

be adjusted to the mean value of the base element. Appendix II This section outlines the procedure that was used to predict from circuit considerations the uncertainty intro- duced in a measurement of the combined effects from 1) the finite resistance of the four-terminal junction, 2) the changes in connections when transferring from a

series to a parallel mode, and 3) the variations in lead and contact resistance as each section is introduced

into the bridge. Since the uncertainties would be greatest

for the lowest valued sections, only the first decade was

considered. Three circuits must be examined. Two of these duplicate

the networks associated with the two principal methods

of measurement while the other identifies the standard

as its ratios appear in use. It is necessary that the ratios measured be identical to those in use or that a correction be applied to compensate for the discrepancy.

A four-terminal junction that departs from the ideal configuration has the equivalent circuit of Fig. 6 [6]. As

applied to the junctions of the DD^l, the six resistances are calculable from the plug resistance of the connectors,

1
DZrCBA AND DUNFEE : RESISTfVE VOLTAGE-RATIO STANDARD \

y©

M a - a Vxy

e b = b'Vx£ r c=c-yxy

(D e=2(x+yxy)

f=2(y+yjy)

x = DIRECT RESISTANCE = Vl2/l s« y= CROSS RESISTANCE =V13/I24 x»y=5.5 ±0.5/i Q

ofeb'ePLUG RESISTANCE OF CONNECTOR cWraLEAD RESISTANCE OF SECTION

Fig. 6. Equivalent circuit of a four-terminal junction.

the lead resistance of the sections and the readily measureable direct and cross resistances X and Y(X Y =

5.5 ± 0.5 iiti). Fig. 7(a) identifies the circuit of DD-4

when it is used in the NBS ratio comparator to measure an unknown ratio. (The 10:1 ratio is indicated in the

figure.)

Making the appropriate A-Y transformations reduces the circuit to that of Fig. 7(b). The ratio in use

N« =

t-i

ft. = Nu [l + cj

where Rk = R'k + Yk + d'k + c'k+1 . The problem now reduces to a similar examination

of the two measuring networks to determine the extent

to which the measured ratios differ from those in use. The measuring network for the bootstrap method is indicated in Fig. 8(a) and simplified in (b), where

<7*(or hk+ i) is the sum of the socket and contact resistances and lk (oT mk+ i) is the lead resistance. Applying the latter network to each successive measure- ment through the decade gives a defining equation of the form

L t"l —J bootitrap

= n\l + £ dk /n - £ fa" + «» + t3t) 1

where

elt = [(X, - X) + (XM - X2 )]l/ft,

«2 t = [(a£ - a'i) + (02 - + (#* - 0i) + Qi2 - hk+1)]l/R,

«3t = [(h - h) + (m2 - mk+l )]l/R l

and

£<**

From an experimental study of the junctions, connectors, and connecting leads, it is unlikely that the sum of the

WV «««. El

y,»d, Rj c2 y2»d2 R2 c^d^yj R's

R'^ cj,

•2 k

(b)

Fig. 7. Circuit of DD-4 (a) and its equivalent (b) when in use.

Fig. 8.

TO BRIDGE

(b)

Circuit and its equivalent for DD-4 when measured by

bootstrap method.

« terms would exceed 0.03 ppm. It appears that the ratio measured and the ratio when used are identical to well within the accuracy assigned.

The circuits for the series-parallel method are shown

in Fig. 9 and refer to the connections when the reference

section and the parallel array of the first decade occupy

the R arm.

All resistances are identified as before except for two

pairs of compensating resistors (u,—u2 ) and {v l —v2 ). Resistors u,. and Vi are adjusted to bring points -p x and p2 to the same potential. Resistors u2 and v2 are adjusted

to bring points p3 and p4 to the same potential. Since

the a leads, in contrast to the v leads, carry current to or from two branches, their resistances are half those 362
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, NOVEMBER 1970

B s 6

(a)

a y&r P3

foi fs by <£ \d5 3Rp

"2

"l PI

v, P2

cKej^/Oe s'2

3Ri

P4

"2 s

i m2

TO BRIDGE

(b)

Fig. 9. Circuits for DD-4 when measured by series-parallel method.

of the v leads. The corresponding simplified circuits are shown in Fig. 10. The balance equation when the first section occupies the R arm is

S[l + D, - D0 ]

where Rt is that portion of the section that is identical to the section when in use and the a/3 terms are the resis- tances of the parallel arrays at the two ends. The balance equation when the parallel group occupies the R arm is

S[l + Dp - D0]

= 4 + H« + "i + ^ + «TT^ + «Tfs)]

From these two equations and making the parallel-toseries conversion, the defining equation for the 10:1

ratio is t|^- = 10[l + |

d, + ^(eJ + ^ + 4i)]

where

e{ = (Xn - 9X2)1/9B,

el = [(« - k) + (ml - m,)]l/fl,

/ = \( _ gjgi

63 ) LVa; + a, + ft/

\a2 + ft a2 + ft/

J

and

10 ~

Cl0 '

The sum of the «' terms is calculated to be about

0.02 ppm. From this value and that given for the « term

of the bootstrap method a difference no greater than

0.05 ppm would be expected between the two methods.

This is in good agreement with the value of 0.02 ppm

deduced from experiment for the 10:1 ratio. Appendix III The -principal argument of the analysis deals with the

results obtained for the 10:1, 100:1, and 1000:1 ratios by the bootstrap and series-parallel methods, since these were more apt to disclose junction and connection

uncertainties as well as uncertainties arising from environmental influences. Results for the intermediate ratios that are available only from the bootstrap method provide

additional information. The supplementary methods

further substantiate the argument.

Data from the first series of measurements obtained

at 50 percent rated voltage for each of the three ratios are analyzed separately from those of similar sets obtained

subsequently at rated voltage. The principal argument

covers data obtained after the system was relocated in a room environment that was more easily controlled. This

required the deletion of only three pairs of measurements

for each ratio. All results are stated as corrections to the nominal

ratios in ppm.

Range

To first provide a feel for the data and the magnitudes

involved, all data, irrespective of elapsed time, method,

or innovation were plotted as histograms (not included)

for each of the ratios at the two voltages. The values

at rated voltage for each ratio approximate a normal

distribution and the same is true for the values at reduced

voltage when a few of the initial and relatively extreme

outliers are deleted. Under this condition, the average

363
DZIUBA AND DUNFEE: RESISTIVE VOLTAGE-RATIO STANDARD

°i +y.

d,+x, -Wr

b'2 +yj

TO BRIDGE

(a)

Fig. 10. Equivalent circuits for DD-4 when measured by series- parallel method. In text, on = ui + si + ai + Y,

ai' = ui + si + at' + Xi

0:2 = v2 + + a2

' + Yi

fii = »i '+ s,' + 6/

fit = «i + *i' + W + Yt

fix = Vi + Si' + fen' +

fit = ui + si + a*' + Xa

TABLE I Ranges for all Measured Values and Their Averages (ppm)

Percent Correction Range

Rated Average (ppm)

Ratio Voltage 71 (ppm) Maximum Minimum

10:1 50 .37 -0.072 +0.13 -0.30

100 59 +0.199 +0.28 +0.08

100:1 50 40 +0.528 +0.96 0.00

100 59 +1.012 +1.26 +0.56

1000:1 50 43 +1.432 +1.89 +0.84

100 53 +1.456 +1.73 +1.14

and range have the values given in Table 1, where n

specifies the number of measurements. Inclusion of the

outliers increases each range at reduced voltage by

about 1 ppm. The corresponding effect on the averages

is no greater than 0.2 ppm.

t Test This portion of the analysis is an examination for any

difference between the two principal methods. A sufficient number (n) of measurement pairs extending over the 4-month period were available for which a value by the one method was obtained within 1 hour of the other. The 99 percent confidence interval estimates of the average difference between the methods are given in Table II along with the averages d and the standard

deviations sd of the differences. The difference was taken

as the bootstrap value less the series-parallel value. The following facts and apparent inconsistencies are

to be noted.

1) There is strong evidence at rated voltage of a negative

difference between the methods that is essentially negligible for the 10:1 ratio but which increases to almost

TABLE II*

Percent Rated

d

Confidence Interval Ratio Voltage n Sd Estimates

10:1 50 13 +0.042 0.089 -0.033 -» +0.117

100 2S -0.008 0.033 -0.025 -» +0.009

100:1 50 13 +0.101 0.082 +0.031 -> +0.171

100 28 -0.047 0.036 -0.066 -> -0.028

1000:1 50 13 +0.056 0.211 -0.123 -» +0.235

100 28 -0.250 0.044 -0.273 -» -0.227

* The 99 percent confidence interval estimates of the average

difference between the two principal methods with the averages and

the standard deviations of the differences included.

0.3 ppm for the 1000:1 ratio, i.e., for the 100:1 and

1000:1 ratios, the limits [d - (tsd /Vn)} < 0 < [d +

(tsd / y/n)] are not satisfied. 2) A consistent set of limits is lacking at 50 percent

rated voltage, and the intervals are wider.

3) The standard deviations at rated voltage are es- sentially equal and small in magnitude but exhibit perhaps a slight upward trend with increase in ratio. 4) The corresponding standard deviations at reduced

voltage are inhomogeneous in magnitude with sd at the 1000:1 ratio being unduly large. The set lacks the self- consistency feature exhibited by that at rated voltage.

The data and associated test conditions warranted

further examination.

Data at Rated Voltage

Further examination of the data and the graphs of

Fig. 11 indicates that d and the confidence interval for the 1000:1 ratio at rated voltage are more negative than

would be expected. Several known factors could account

for the apparent difference between methods, but these

are equally applicable at the lower ratios and at reduced

364
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, NOVEMBER 1970

Fig. 11. Corrections to the ratios at rated voltage in ppm versus elapsed time. Solid line indicates bootstrap method; dashed line indicates series-parallel; arrows indicate desiccant was changed after the given measurement.

(a) 10:1 ratio, (b) 100:1 ratio, (c) 1000:1 ratio.

voltage, with one notable exception. The power dissipated

in the resistance elements in the bootstrap technique

is nine times that in the series-parallel, and any contribution to the difference from this source would be magnified

at the higher voltage.

The temperature coefficient of ratio is negative for

all ratios and attains, with the bootstrap technique, a maximum of —0.6 ppm/°C for the 1000:1 ratio. Consideration of the power per centimeter squared of radiating

surface with the resistive elements in contact with the

circulating air leads to an estimated temperature rise at the surface of about 0.3°C at rated voltage. This

would account for an equivalent change in the 1000:1

ratio of about —0.17 ppm. It is reasonable to assume a

negligible heating effect for the parallel mode. With

this assumption, it would be expected that values by

the bootstrap method for the 1000 : 1 ratio would be more negative by the above amount than those by the

series-parallel method.

If the bootstrap values are adjusted upwards (or

series-parallel downwards) to eliminate the difference

arising from heating effects, the confidence interval becomes (-0.103 -» -0.057) in contrast to (-0.273 —> -

0.227) given in the table. A similar treatment of the data

for the 100:1 ratio reduces the limiting values given in the table by about 0.010. Thus, the evidence supplied

by the data at rated voltage would indicate a real difference approaching 0.1 ppm between the two methods.

Data at Reduced Voltage

An equivalent concinnity among the data at reduced

voltage would be expected unless spurious influences, not present to the same extent at rated voltage, were

affecting the measurement. Two obvious factors must

be considered—an inherent instability in the ratio stan- dard and effects from humidity. Since the series of measurements at reduced voltage preceded those at rated

voltage, either or both might exist. However, since the two series were immediately consecutive, an inherent

instability in the standard appears a less likely cause. 365
DZIUBA AND DUNFEE: RESISTIVE VOLTAGE-RATIO STANDARD 277

Furthermore, with self-heating greatly reduced, changes

in the resistance of the elements would be reflected in both methods unless the instability were of an unusual

nature, e.g., a steady but rapid drift. No evidence of

this behavior existed. It is more logical to suppose that a humidity not under careful control would be the more

likely cause for the discrepancies in Table II, since its effect on the response of resistive networks is difficult to predict. This conclusion is reached in the next section. Table III completes the present argument and intro- duces the section that follows. It includes the standard

deviations sb and sp for the bootstrap and series-parallel

methods, respectively, and the ratios of their squares

F. The values at rated voltage are included for comparison.

There is no evidence from the F values that the precision

differs for the two methods at either applied voltage.

As indicated later, it is probable that the poorer agree- ment between the standard deviation pairs sk and s„ at reduced voltage arises from insidious humidity effects.

Humidity Influence No attempt was made to control the humidity within

the enclosure during the measurement series at reduced

voltage. The small effects from this parameter were

given attention during the measurement series at rated voltage when it appeared that an accuracy of 0.2 ppm

or better was achievable. During the latter period, changes were effected in the humidity condition by replenishing a drying agent

at suitable intervals. These changes are entered in Fig. 11 where the arrows along the abscissa indicate that the drying agent was replaced after the corresponding reading on the ordinate. The humidity effect is evident in all cases, and no trend with elapsed time of any significance

is apparent. Changes in ratio following replacement of the desiccant are in the same direction for the two methods

and agree to 0.1 ppm or better. An examination of values obtained for all nominal

ratios by the bootstrap method disclose with one exception

the following behavior.

1) On the XI range, corrections for all ratios go more

negative with a maximum at 0.2 ppm.

2) On the Z10 and X100 ranges, corrections for all ratios go more positive with maximum at 0.3 and 0.2 ppm, respectively.

3) As to the exception and for no apparent reason, the signs of the changes are reversed to those given

in 1) and 2). In contrast to Fig. 11, similar curves for the data at reduced voltage (omitted from the paper) exhibit downward trends for the two highest ratios. The maximum

TABLE III Individual Standard Deviations for the Two Principal

Methods and the Ratio of the Variances Squared

50 Percent Rated Voltage Rated Voltage

Ratio n Sb sp F n Sb sp F

10:1 13 0.093 0.062 2.25 28 0.048 0.045 1 .15

100:1 13 0.151 0.123 1.51 28 0.134 0.134 1 .00 1000:1 13 0.100 0.165 0.37 28 0.110 0.107 1 .05

apparent drift amounts to about 0.3 ppm for the seriesparallel method on the 1000:1 ratio and about 0.2 ppm

for all others. Furthermore, the effects from the first recorded change in the drying agent, made near the end of the series, are appreciable. The changes in values

for the respective methods differ by 0.2 ppm but have

a maxima of 0.6 and 0.4 ppm for the 100:1 and 1000:1

ratios. Omission of the data at reduced voltage in assigning an uncertainty of measurement is based on the facts and argument presented. These indicate that the measure- ment process was not under statistical control during

operation at reduced voltage.

On the basis that the relative humidity is now closely

controlled, the standard deviation a is taken as the average of sp and sb as given for the 10 : 1 ratio. The uncertainty of measurement is then estimated at 0.2 ppm,

based on the 3c confidence limit for random errors, and 0.1 ppm for the systematic error. Occasional values obtained by the two "check" methods substantiate the above conclusion.

References

[1] B. L. Dunfee, "Method for calibrating a standard volt box,"

J. Res. Nat. Bur. Stand., Sec. C, vol. 67, pp. 1-13, January-March

1962.

[2] R. Ohlon, "A new type of volt box," Metrologia, vol. 5, pp. 21-25, 1969.

[3] "Metrology catalog," Electro Scientific Industries, Inc., Portland, Ore., 1970-1971.

[4] F. K. Harris, Electrical Measurements. New York: Wiley,

1952, p. 272.

[5] F. Wenner, "Methods, apparatus and procedures for the comparison of precision standard resistors," J. Res. Nat. Bur. Stand., vol. 25, pp. 279-285, August 1940.

[6] J. C. Riley, "The accuracy of series and parallel connections

of four-terminal resistors," 1965 IEEE Internal. Conv. Rec,

pt. 2, vol. 13, pp. 136-146.

[7] C. H. Page, "Errors in the series-parallel buildup of four- terminal resistors," Res. Nat. Bur. Stand., Sec. C, vol. 69, pp. 181-189, July-September 1965.

[8] F. B. Silsbee, "A shielded resistor for voltage transformer

testing," Sci. Papers Nat. Bur. Stand., vol. 20, pp. 489-514, December 1925.

[9] "AC calibration—A systems concept," Holt Instrument

Laboratories, Oconto, Wis., March 1962.

[10] J. J. Hill, "Calibration of dc resistance standards and voltageratio boxes by an ac method," Proc. Inst. Elec. Eng., (London)

vol. 122, pp. 211-217, January 1965.

366
JOURNAL OF RESEARCH of the National Bureau of Standards—A. Physics and Chemistry

Vol. 65A, No. 3, May-June 1961 Evaluation of the NBS Unit of Resistance Based on a

Computable Capacitor

Robert D. Cutkosky

(January 16, 1961)

An evaluation of the unit of resistance maintained at the National Bureau of Standards,

based on the prototype standards of length and time, is described. The evaluation is based on a nominally one-picofarad capacitor whose value may be calculated from its mechanical

dimensions to high accuracy. This capacitor is used to calibrate an 0.01 -microfarad capacitor. A frequency-dependent bridge involving this capacitor establishes the value of a

104 -ohm resistor. Comparison of that resistor with the bank of one-ohm resistors maintaining the NBS unit of resistance establishes that this unit is

fisc=1.000002 3 ohms ±2.1 ppm. The indicated uncertainty is an estimated 50 percent error of the reported value based on

the statistical uncertainty of the measurements and allowing for known sources of possible

systematic errors other than in the speed of light, assuming that the speed of light

c = 2.997925X1010cm/sec.

1. Introduction

The United States national standard of resistance

is maintained at the National Bureau of Standards

by a group of 1-ohm manganin resistors. Values of

resistance are assigned to the 1-ohm reference standards by techniques based ultimately upon the

national prototype standards of length and time.

The most accurate methods which have been used

for this purpose in the past made use of either self- inductors or mutual inductors whose inductances were computable from their mechanical dimensions. In principle, it is then straightforward to compare the

impedance of the computable inductor at a known

frequency with that of the 1-ohm reference standard resistors. In practice, this comparison is likely to involve several steps, and may not even make use

of sinusoidal currents through the inductor.

An alternative method for assigning values to the

reference resistors involves constructing a capacitor

whose capacitance may be calculated from its mechanical dimensions. By comparing the impedance of

the capacitor at a known frequency with that of the

reference standard resistors one may establish the

values of the reference resistors in terms of the

prototype standards.

Within recent years the development of an improved computable capacitor [1, 2, 3, 4] 1 and improved methods for the precise comparison of

capacitances [5, 6] have made the latter process

appear to be the more fruitful. The method involves stepping up from a 1-pf

three-terminal computable capacitor to two 0.01-/xf three-terminal capacitors. The step-up is made in

: Figures in brackets indicate the literature references at the end of this paper. 585402—61 2

four steps, using a transformer whose nominally 10:1

ratio may be accurately measured. The average

admittance of the two O.Ol-^f capacitors at an angular frequency of 104 radians per second is compared

with the average admittance of two 104-ohm shielded

a-c resistors using a bridge network to be described.

These measurements serve to assign values to the

104-ohm a-c resistors in terms of the prototype

standards.

A conventional d-c step-down from these 104-ohm

a-c resistors to the 1-ohm NBS reference resistors

provides an absolute calibration of the 1-ohm resistors. A small correction must be applied to the

results so obtained because of the a-c-d-c differences

of the 104-ohm resistors. This correction is determined by comparison with a special transfer

resistor.

2. Computable Cross Capacitor

Computable capacitors of the traditional guardring or guard-cylinder design require, in general, the

measurement of several dimensions. For example,

with the guard-ring type, area, plate separation,

flatness of electrodes, parallelism, aod proper location

of the guarded island are of major concern. Most of

these dimensions have to be measured to a higher

accuracy than the required overall accuracy of the

computed capacitance.

The development by A. M. Thompson and D. G.

Lampard of a device called by them a computable

cross capacitor [1] has made possible the construction

of a capacitor whose value may be computed from

dimensional measurements with an accuracy considerably higher than that attainable with any

previous design.

367
A cross capacitor consists of a cylindrical arrangement of four electrodes, with, cross section as shown

in figure 1. Electrodes 1, 2, 3, and 4 are separated

from each other by very small insulating gaps. Let

Ci/L be the direct capacitance per unit length (perpendicular to the plane of the drawing) between

electrodes 1 and 3 with electrodes 2 and 4 grounded,

and let &./L be the direct capacitance per unit length

between electrodes 2 and 4 with electrodes 1 and 3 grounded. It may be shown that in cgs, esu,

g-4*2Cl/£_|_ g -4r2C2 /L =lj0r m ^{JvS UllitS,

e,L(hi2/ir)[l +0.087 (AW)2

]

+ <C'{(AC/Cy}] farads (1)

where e 0 =l/n0c

2

, c is the speed of light, and AC—

C2 — Cx

. The notation means that all remaining

terms in eq (1) are fourth order or higher in AC/C,

and their sum approaches zero as K(AC/C)* as AC/C tends to zero. This relationship allows C

to be calaculated from a single measured length L. One finds that if C^d, then C'/L«2 pf/meter«

0.05 pf/in.

The requirements that both C and L be easily

measurable have led to the use at NBS of two„ cross capacitors in parallel, each about 10 in. long, to produce a capacitance of about lpf [6]. Figure 2

shows the cross section of the NBS 1-pf computable

cross capacitor, which is housed in an evacuated

chamber. Electrodes 1, 2, 5, and 6; and electrodes

2, 3, 6, and 7 of the two cross capacitors are steel bars

% in. in diameter supported with accurately ground

glass slips and disks, shown dotted in the figure,

resting on a precision granite surface plate. The

bottom row of bars is held together with a phosphorFigure 1. Cross section of a general cross capacitor.

bronze coil spring between bars 4 and 8 at each end

of the assembly. The glass spacers are all located

within guard sections at the ends of the capacitor,

so that no solid dielectric appears between the bars

within the active length of the capacitor. Figure 3

shows a side view of electrode 6, which is similar to electrode 2. The assembly is constructed of precision gage blocks having circular cross sections,

with wrung joints between t lie 10-in. bar and its adjacent 2-in. bars, and insulated witli firmly

cemented 0.002-in. mica spacers between each pair

of 2-in. bars. The outer 2-in. bars constitute the

guard sections, and are grounded. The composite

14-in. central section defines the length of the cross

capacitor, and extends roughly between the centers

of each insulated gap. Electrodes 1, 3, 5, and 7 are approximately IS in. long, and so extend well beyond

the active sections of bars 2 and 6. Additional shields not shown in figure 2 eliminate

capaeitive coupling between electrodes around the

outside of the grounded electrodes. This is essential,

since (1) deals only with the capacitance associated

with the region in the interior of the electrode system.

One capacitance measurement is made, with elec- trodes 1, 3, and 6 grounded, of the direct capacitance

between electrodes 5 and 7 in parallel and the central

section of electrode 2. Another measurement is made, with electrodes 5, 7, and 2 grounded, of the

capacitance between electrodes 1 and 3 in parallel

and the central section of electrode 6. If we neglect,

for the moment, the small second order correction

0.0S7(AC/6') 2 in eq 1, the average of these two

capacitances is equal to 2X e 0 LTn2/V; where L' is the

average length of the two central sections of electrodes

2 and 6, or approximately 14 in. If the 10-in. gage

blocks are removed from electrodes 2 and 6 and the 2-

in. bars on either end are wrung directly together, a new cross capacitor is formed with a length differing

'/////////// / / / / /

/ / / / / /

/

/ ///////

'//////////'// ///////// / s ' ' ' '/////A'/''''''''/' //x /

/

/ //// ' / / / / / '// '// /granite surface plate / // ///// / ^, ///,//////// '////'/////

//////// / // ''/////// //////////

/////// / / / ////// / / / ////// / / / /

/

Figure 2. Cross section of the NBS cross capacitor.

«— .002" .002" —

Figure 3. Side view of the XBS cross capacitor detector

electrodes.

368
from the first by twice the average length of the 10-in. bars, or about 20 in. The difference between the

two cross capacitances so measured is not affected

by irregularities or lack of perfect centering of the

bars near the insulated gap, provided that the end

sections are not rotated or otherwise disturbed Avhen

the 10-in. bars are removed or replaced.

The small discontinuities caused by the slightlyrounded ends of the gage blocks produce only secondorder errors in the capacitance computed from eq

(1). These errors have been investigated experimentally by introducing additional discontinuities on the bars.

Between May 12 and May IS, 1960, and again

between July 21 and July 27, 1960, two series of

measurements with the NBS computable cross capacitors were made to establish the value of a

fixed, 1-pf, air-dielectric, three-terminal, reference

capacitor; NBS S9790-B. This capacitor was maintained in an oil bath at a constant temperature near

25 °C.

A transformer ratio-arm bridge was used for this purpose with the circuit shown in figure 4. The

cross capacitor with the 10-in. bars in place has a capacitance slightly under 1.4 pf so that the bridge

may be balanced with a small adjustable capacitance

p as shown, and a small adjustable conductance not

shown, which may be placed on either side of the

bridge. The bridge components have been described

in detail elsewhere [6] and "will not be elaborated

upon here.

The bridge circuit for the measurement with the

10-in. bars removed is similar to figure 4 except

that the cross capacitance is now about 0.4 pf, so the 1-pf capacitor in the lower portion of the circuit

must be removed. The 4-pf capacitor consists of

4 one-pf capacitors which may each be compared

directly with the 1-pf standard, and then connected

in parallel. Since the adjustable capacitor has a

relatively high reading, it is also necessary to calibrate

it for each measurement.

CROSS

I CAPACITOR

All of the above steps are straightforward and can

be done in a symmetrical fashion (calibration, cross- capacitor measurement, and recalibration) in less than 30 min by an experienced operator.

Before and after the determination, the 1-pf

capacitor used is compared with the 1-pf reference

capacitor in the oil bath.

Although small direct (three-terminal) capaci- tances are much easier to measure than small twoterminal capacitances because of the virtual elimination of connection problems, a particularly trouble- some source of error arises when capacitance to ground is very high, as is the case with the NBS

cross capacitor. Figure 5 shows a three-terminal

capacitor with direct capacitance Ci connected to a bridge with leads of inductance Lu L2 , and L3

. It may be shown that the effective capacitance seen by

the bridge is given by

Ce//~Ci[l

o>

2 (C2 C3 ] L\ + w

2 C3 Ln 4- or Co T-i'i 4-

For Ci, C2 , and C3 all less than 1000 pf, the important

error, expressed as an additive term, is — orLiCoC3 ; which is independent of Ci and hence relatively more

important for a very small C\. Thompson [5] has

shown that if the capacitor is treated as a fourterminal network as in figure 5 with a high permeFigure 4. Inductively coupled ratio arm bridge used with the XBS cross capacitor.

Figure 5. Three-terminal capacitor G connected to a bridge with three leads, with four leads and using a high permeability

core (See text).

369
ability core linking one ground lead with one active

electrode lead, the troublesome error term is eliminated. This technique has been used throughout

the measurements reported here. It is impossible

with the NBS cross capacitor to completely eliminate

all common ground lead inductance in this way, and

a small inductance L\ remains in the cross capacitor

circuit. We have found it necessary to measure Ctfs as a function of frequency and extrapolate to zero frequency under the assumption that the error

is proportional to w

2

. With most of L\ eliminated by

the use of high permeability cores, the factor

(JLiC^Cz is very small, and the d-c value may be

determined to high accuracy.

Table 1 shows the results of two complete series of

measurements on the NBS computable cross capacitor, based on the assumption that the reference

capacitor in the oil bath was exactly 1 pf.

The correction term in the measured capacitance

0.087 (AC/C) 2 is less than two parts in 107 and has been

neglected. The calculated cross capacitance also appears in table 1, based on the measured lengths of

the 10-in. gage blocks, and taking the speed of light

c=2.997925X1010 cm/sec as adopted recently by the

IGGU and the URSI. Combining these figures we

obtain true values for the reference capacitor at the

time of measurement. The difference between the

May and July figures reflects a shift in the temperature of the oil bath which occurred some time in

June, and has no effect on the final results quoted in

Table 1. Capacitance of cross capacitor

Based on 1-pf reference capacitor #NBS 89790-B.

Time and date (1960) Nominal

length Frequency Measuredcapacitance D-c capacitance

in. cfs Pf Pf

11:00 a.m. May 12 14 1592 1. 3898231 1. 3898241

8:55 a.m. May 13 14 1592 1.3898224 1.3898234

9:40 a.m. May 13 14 1592 1. 3898222 1.3898231

2.05 p.m. Mav 13 14 1000 1. 3898229 1. 3898233

2:50 p.m. May 13 14 1592 1.3898220 1. 3898229

1:20 p.m. Mav 16 4 1692 0. 3973528 0. 3973532

2:00 p.m. Mav 16 4 1592 . 3973531 . 3973535

2:55 p.m. Mav 16 4 1000 . 3973536 . 3973538

10:45 a.m. May 18 14 1592 1. 3898201 1. 3898210

11:35 a.m. Mav 18.... 14 1000 1.3898211 1.3898215

12:20 p.m. May 18.... 14 1592 1.3898198 1.3898208

Measured cross capacitance difTerence=0.9924687 pf. Computed cross capacitance din"erenee=0.9924151 pf. True value of reference capaci tor = 1-pf —54.0 ppm (May 1960).

Based on 1-pf reference capacitor #\T BS 89790-B.

Time and date (1960) Nominal

length Frequency Measured

capacitance D-c capacitance

in. c/s Pf Pf

9:22 a.m. Julv 21 14 1592 1.3898177 1.3898187

10:15 a.m. July 21 14 1000 1.3898177 1.3898180

12:45 p.m. Julv 21 14 1000 1.3898171 1.3898175

1:15 p.m. Julv 21 14 1592 1.3898164 1.3898174

9:45 a.m. Julv 25 4 1592 0. 3973511 0.3973515

11:20 a.m. Julv 25 4 1000 .3973511 . 3973513

1:00 p.m. Julv 25 4 1000 . 3973512 .3973514

1:43 p.m. Julv 25 4 1592 . 3973507 .3973511

9:15 a.m. Julv 27 14 1592 1.3898154 1. 3898164

10:05 a.m. Julv 27 14 1000 1.3898165 1. 3898168

12:52 p.m. Jufv 27 14 1000 1.3898155 1. 3898169

1:15 p.m. July 27 14 1592 1. 3898165 1.3898175

Measured cross capacitance difference=0.9924661. Computed cross capacitance din"erence=0.9924151. True value of reference capacitor=l-pf -51.4 ppm (July 1960).

this paper. The stability of the reference capacitor

and the closeness of temperature regulation during

each of the determinations was sufficient to reduce

any error arising from the variability of the reference

capacitor considerably below 1 ppm.

3. Capacitance Step-Up

The bridge for comparing capacitive reactance with

resistance, to be described in a later section, requires

the use of two 0.01-Mf capacitors. These capacitors

must be calibrated in terms of the 1-pf reference capacitor. A transformer ratio-arm bridge with a nominally 10:1 ratio is used for this calibration in the

circuit of figure 6. The balance conditions are, to a sufficiently close

approximation,

C? =10 Ca + a)-?, and g2 =lO g^-lOuCrf-q,

where p and q are small adjustable admittances for

balancing the bridge. One may connect p and q to point ' B" instead of "A" to change their apparent

signs if this is needed to obtain a balance.

Shielding completely surrounds each of the ca- pacitors in the circuit in such a way that the measure- ment compares the direct capacitances Ci and C2 . This technique allows the direct capacitance of the

10-pf capacitor to be determined relative to the 1-pf

reference capacitor. The 10-pf capacitor is in turn

compared with a 100-pf capacitor, to reach after four

such steps the 0.01-juf mica capacitors used in the

resistance bridge. All of the step-up capacitors are located in a regulated oil bath. The 0.01-Mf ca- pacitors as well as the 1000-pf capacitor are suffi- ciently large to require correction for the equivalent

series inductances of the transformer. The total in- ductance correction was found to be two parts in 107

for each 0.01-juf capacitor [6]. It was desired that the total uncertainty in the

step-up be less than one part in 107

, which requires

that the transformer ratio l0(l + a+j/3) be known to

e 2

= - ioei (

i

a + j 0 )~ioov

e,=iov Figure 6. Ten-to-one inductively-coupled ratio arm bridge for stepping up from 1 pf to 0.01 nf in four steps.

370
an accuracy of about two parts in 10s

. A procedure

for calibrating such a transformer to the required

accuracy has been described [7] which involves the

use of 11 nominally equal three-terminal capacitors

having their detector electrodes connected together.

Provision is made for switching any one of the ca- pacitors to the high voltage side of the transformer

while the other 10 are in parallel on the low voltage

side. The bridge was balanced with a small admittance on one side or the other of the bridge with each

of the 11 capacitors in turn on the high voltage trans- former winding. After applying small corrections for lead impedances one obtains from this series of measurements values for a and /3 in the expression for the

transformer ratio.

4. Resistance-Capacitance Bridge

Several bridges capable of being used for the comparison of resistance with capacitive reactance have

been used in the past, one of the most popular being

the Wien bridge. The Wien bridge, like many fre- quency dependent bridges, is difficult to shield

against stra3r pickup. The difficulty comes from the

fact that one arm of the bridge contains a resistor and

a capacitor in series. If this arm is shielded, leakage

currents from their common point to the shield produce errors which are difficult to evaluate with high

accuracy.

4.1. Bridge Equations

In theory it is simple to compare resistance with

capacitive reactance provided that one may obtain

two a-c voltage sources exactly 90° out of phase and

exactly equal in magnitude. Figure 7 shows two such

sources used to compare a capacitance C with a con- ductance G. The balance condition is <?=wC. The

components may be readily shielded as shown with

dotted lines; leakage paths to ground are either across the detector, which at balance has no voltage on it; or across one of the generators, which is assumed to have a very low impedance. Although it has not been

possible to construct voltage sources with the re- quired phase and magnitude relationships, accurate

low impedance voltage sources equal in magnitude

and 180° apart in phase are readily procurable from

a center-tapped transformer.

A double bridge using the principles of figure 7 may

be constructed with the circuit of figure 8, which

shows all voltages referred to one generator as refer- ence.2 The voltages +1 and — 1 (1 +a+j0) are approximately equal and 180° out of phase, with a and

@ representing magnitude and phase angle errors re- spectively. The voltage j (1 + z) is approximately 90°

out of phase with each of the other generators. The

small complex number z represents the departure of

the generator from its nominal value.

Figure 8 also shows the residual leakage conductances gx and g2 of the main capacitors Cx and C2 , and

the stray capacitances c x and c2 of the main conductance standards Gx and G2 .

1 Bridge networks of this type were investigated at the National Standards Laboratory in Sidney. Australia, in 1952 but the findings were not published. The network described in this paper was subsequently developed independently.

Figure 7. Basic bridge for comparing conductance with capacitive susceptance.

Figure 8. Double bridge for comparing conductance with capacitive susceptance.

This circuit is used with Ci — C^—O.Ol p.{, Gi~

G2 ~l0~i mhos, and w«104 radians per second

(about 1592 c/s). With both bridges balanced we have the conditions

and

jwc2 +G2 =—j(l + z) (jwC2 +

g

2 )

( 1 + a +jp) (juCt+gJ =j(l + z) (j«c, +Gi).

Eliminating z from these equations we have

(jac2 +G2) (jwCi +Gi) = - (1 +a+ilS) (jo>C2 +g2 )(juC1 +g1 ). Separating real and imaginary parts we find

GiG2 =(l + a) tfdC2 +/3 (uCi02

+uC2gx ) — (1 +a)glg2 +u?c1 c2,

371
and

«[c, G2 +CtGA + ( 1 + a) + agi ] = (Su'd C2 -Pgl02 .

It may be shown that if all residual parameters are

small, we may write approximately

G= (l+i a) w£ (2) and

ucG+uC^hp^C- (3) where bars over letters signify the average of the two

values involved.

The voltage appearing at the terminals of detector

1 may be reduced to zero by making either C\ or Gi

and either c t or 17, adjustable, and similarly for detector 2. One finds that even if z is zero, either Ci or

<7i and either c2 or g3 must be negative. As shown in figure 9, this is accomplished by connecting an

adjustable conductance q: from point A to Z), and an

adjustable capacitance pi from point B to D2 . The

main capacitors C\ and C2 consist of the 0.01-juf

mica capacitors in parallel with adjustable capacitors

having ranges of 1 pf. With these provisions eq (3)

is replaced by

co^+WCy-K^1 t? +a)C2 1 ) = ^o;2 C

T2

. (4)

It is apparent from figure 9 and the above equations that C2 is the effective capacitance including

the correction resulting from self inductance between

junctions C and D2 , and Gx must include the series resistance correction between junctions C and

If the open circuit transformer-ratio parameters a and /3 are used, Cx and G2 must include the effects of impedances ^ and f2 , measured from junctions Dx and D2 respectively, through the transformer to the

ground point G. To avoid the problems of ac- curately measuring the transformer impedances, we

have found it profitable to measure the loaded rather

than the open circuit transformer ratio between A

Figure 9. Complete bridge with auxiliary components.

and B to G while the bridge is balanced. Using the

values so obtained for a and j3' , the required values

of Go and d include impedances onlv to junctions

.4 and B.

The loaded transformer ratio is easily measured

with a third bridge involving C3 , (\, and detector 3 on the left of figure 9. This bridge is balanced with

the circuit as shown and rebalanced after interchanging C\ and (\. The readings of p2 and <£> are

called i> 2 and q-2 before interchanging, and p2 " and

q2 " after interchanging. The transformer parameters a' and /3' are calculated from the formulas

P

co(C?3 -f-C4 ) The signs of p2 and q2 are positive if connected to B

as shown, and negative if connected to ^4. It is found that for this bridge, with the loading shown

in figure 9, a'~ + 17XlCT6 and p' = - 16X 10-6

.

4.2. Physical Bridge Arrangement

In order to minimize current loops and localize

the junction points ^4, D2 , C, Du B, and D3 , a hexagonal bridge center was constructed of copper as shown in figure 10. Each segment consists of five

coaxial connectors with their shields bolted to a

%-ia. copper top plate and their inner conductors

connected by means of a copper plate. These six connecting plates shown dotted in the figure, are shielded from each other so that capacitance and

conductance between segments arises only from

external elements that are connected between pairs

of coaxial connectors in the various segments. The

intersections of the arms of the copper connecting

plates define the junction points referred to in the

discussion of figure 9. The bridge center is located

just above the liquid level in the middle of a large

temperature-regulated oil bath.

Figure 10. Diagram of bridge center.

372
The voltage sources are connected to the bridge

center points A, B, and C, by heavy wires passing

through a tube that is welded to the bottom of

the oil bath tank and extends above the liquid

level beneath the center. The I04-ohm resistors and

the 0.01-Mf capacitors, hermetically sealed and immersed in the od to maintain them at a constant

temperature, are connected to the center with rigid

coaxial leads positioned as in figure 10. Coaxial detector leads run from the connectors

at Du D2 , and D3 , upwards through an eye-level

shelf to filters, amplifiers, and visual detectors.

Figure 11. Photograph of components shown in figure 10.

Figure 12. Photograph of bridge assembly for comparing con- ductance with capacitive susceplance.

Figure 11 shows the principal features of the bridge

arrangement, and figure 12 shows the auxiliary

admittances in place for balancing the bridge.

Although the two voltage sources derived from

a transformer are easily procurable, the third source 90° out of phase with them presented some

design problems. Our equipment uses a passive

phase-shifting network driven by a separate winding on the main transformer to produce this voltage.

The phase-shifting network is adjustable to allow

the complex components of z to be varied in small

increments and set within a few parts in a million

of zero. The assumptions leading to the bridge equations

require only that the proper voltage at junction

C be produced and maintained, and place no restriction on the equivalent series impedance of

the 90° source. However, if its impedance is too

large, variation of one balancing component such

as d in figure 9 changes not only the voltage at

Di, but also the voltage at junction C and hence

at D2. This mutual dependence between the two

bridges makes it very difficult to balance both

bridges simultaneously unless the impedance of

the 90° source is less than Xo the impedance of the

bridge arms, or in this case less than 1000 ohms.

Based on these considerations, the circuit of figure

13 was chosen to provide the three sources for the

bridge. With A, B, and C connected to the bridge,

the voltage at C may be adjusted, with the controls

provided, to the proper value.

Power for driving the bridge is provided by a commercial tuning fork oscillator driving a power

amplifier as shown in figure 14. The filter is designed

to reduce harmonic content of the signal. The

tuning fork frequency is adjustable in a small

range around a; = 104

.

4.3. Frequency Measurement

The frequency is measured with the circuit of

figure 15. The pulse former preceding the preset

counter proper is part of the preset counter assembly.

The sharp pulse produced therein goes through the

gate when the preset counter registers the desired

Figure 13. Bridge voltage sources.

373
TUNING

FORK OSCILLATORFILTER POWER

audi i ci r p

1

TO BRIDGE TRANSFORMER

< r TO FREQUENCY

MEASURING EQUIPMENT

Figure 14. Bridge power supply.

PULSE FORMER

PRESET COUNTER

TIME INTERVAL COUNTER

STANOARO

100 kc /s SIGNAL Figure 15. Frequency measuring system.

number of counts, and the time interval counter measures the time between two such pulses. The

preset counter is automatically reset after 104 cycles at the input. For u=104 the time interval between

pulses reaching the time interval counter is 2t sec, during which time a 10-Mc/s signal derived from the

100-kc/s standard frequency source is counted and

displayed by the time interval counter. The sys- tem may be checked for internal consistency by using

it to measure 104 periods of a standard 1000-cycle

signal. This equipment monitors the tuning fork

output while the bridge is in use, and allows the

frequency to be measured with an accuracy better

than 5 parts in 108

.

4.4. Detectors

Since the bridge balance is strongly frequency

dependent, the harmonics of the driving frequency

may have relatively high signal levels at the detector

inputs. Sharp tuning of the detectors is not suffi- cient to eliminate trouble from this source, since

two or more harmonics may mix in the nonlinear

first stages of the amplifiers to produce apparent

fundamental frequency signals [8]. Figure 16 shows the circuit of filters which precede

the electronic amplifiers used with detectors Dx and

D2 . With the exception of the input transformer, all inductors are 0.1-henry commercial dust-core toroidal

inductors. The input transformer must have a very

high Q to preserve high signal strength, and was

constructed from a similar inductor by removing a few turns and adding a primary; the turns ratio

being chosen to give a good impedance match

between the bridge and the filter. The filter forms

a bridge which is balanced at the second harmonic

ioo kn

VVV AMPLtFL

Figure 16. Filters for harmonic rejection.

by adjusting Cx so that Ci and Lx resonate, and by

adjusting Rlm Also, C2 and R2 are adjusted to produce balance at the third harmonic. Tuning the

input transformer with C4 and the output circuit

with C3 to the fundamental completes the adjustment of the doubly-tuned filter which attenuates

harmonics higher than the third sufficiently for our

purposes. The filters are housed in boxes constructed of Yi6 in. sheet steel for magnetic shielding.

It has been found that with these filters, second

and third harmonic signal levels are attenuated 140

db with respect to the fundamental, and all higher

harmonics by at least 95 db. Experimentation with

filter orientation shows that stray pickup to the

filters cannot produce an error as large as one part

in 107

. The effectiveness of the filters has been investigated by greatly increasing the harmonic content of

the power signals reaching the bridge. For this

purpose a eliode rectifier was placed between the

tuning-fork filter and the power amplifier. Harmonic levels at the bridge center were measured with

and without this diode. It was found that although

the levels of all measured harmonics were greatly

increased, the bridge balance change was less than

one part in 107. Following the detector filters, commercial tuned

amplifiers are used, which drive the vertical deflec- tion plates of cathode-ray tubes. The horizontal

plates of the cathode-ray tubes are driven by constant amplitude sinusoidal voltages derived from the

tuning-fork oscillator but passing through variable

phase- shifters. At balance a straight horizontal

line appears on the cathode-ray tube, and the phase

shifter may be adjusted so that an unbalance in a main component (Ci or C2 ) opens the line into an

ellipse, and an unbalance in phase (gj or^i) tilts the

line. This provides a phase sensitive detection

scheme which allows the bridge to be balanced with

a minimum of time consumed. The detector sensitivity is sufficient to observe unbalances of one part

in 107 in magnitude or 10~7 radians in phase angle.

4.5. Resistance Comparisons

The bridge system described allows the mean of

two a-c conductances Gx and G2 to be measured in

terms of a measured frequency and two capacitors

Ci and C2 whose values are obtained from a computable capacitor. The conductance standards Gx and

374
G2 each consists of a commercial 104 -ohm woven-wire

resistor mounted in a shielded hermetically-sealed

box and placed in the oil bath. The reciprocal of their measured conductance values is a resistance,

which must be stepped down and compared with

the bank of 1-ohm reference resistors which maintain the unit of resistance [9]. The conductance standards Gx and G2 are completely shielded, and are measured in the a-c bridge

as direct conductances. In section 5 we will describe measurements performed to evaluate the

difference between their conductances as measured

with 1592 c/s alternating current, and their con- ductances as measured with direct current. This

difference may be expected to be small, and will certainly remain constant over long periods of time. We will, at this time, tentatively assume

that the difference is zero, and apply a correction

later. The d-c direct conductances of Gx and G2 are compared with the resistance of a stable 104 -ohm

d-c resistor in the circuit of figure 17, which shows

the provision for eliminating the effect of leakage

resistances R3 and i?4 to the shield. The bridge

is balanced first with point (1) grounded, which

shorts i?4 and places R3 in parallel with Gx ; and

then with point (2) grounded, which places R^

across the battery and places R3 across the 104-ohm

standard i?#505. Neither balance depends upon

i?4 , and a large R3 affects the two balances equally

with opposite sign; hence the average balance gives

the ratio 3

r3 +i?#505 Ai

Interchanging the link and standard, moving the

galvanometer to Dh and repeating the two balances

gives

r4 +fl#505 ^£

ra+Wi X

so that to the accuracy desired 7?#505 = l/6?i(X

[l-\-(Al —A2 )/2B], which allows the resistance of

Gi to be compared with the resistance of the d-c

standard. The measurement is made with Gx connected to the bridge center, and measurements

are made between the same junctions used in the

a-c measurement. A similar measurement of G2 allows the 104 -ohm d-c standard to be measured in

terms of G=V2 (Gi-\-G2 ), which is obtained from the

a-c bridge. The measurement method involves the

well-known double substitution technique with

provision for eliminating the errors which might

be caused by leakage resistance.

The variable arm A is constructed from a 100-ohm

direct reading ratio set with a smallest step of one ppm, but is connected in series with a 900-ohm resistor to give a smallest step of one part in 107

.

'This relationship Is actually an approximation, and depends upon R.i being

large compared with Since the two values of Ai differed in our measure- ments by only a few parts per million, the approximation introduces no signifi- cant error.

(LARGE) rMA

'-

n

G,(l04

ft)

Figure 17. Circuit for three-terminal d-c resistance measurement.

This 900-ohm resistor, a 103 -ohm resistor B, the

104-ohm standard, and the link rest on a mercury

stand in the oil bath.

Table 2 shows the results of several comparisons

of the 104 -ohm d-c resistor with the 1-pf reference

capacitor, using the frequency-dependent bridge.

Recalling that the reference capacitor was measured

in terms of the computable capacitor in May and

July 1960, it will be noted that groups of runs were made immediately preceding and immediately fol- lowing each cross capacitor measurement.

Each set of one cross capacitor measurement and

two groups of resistance-capacitance bridge runs

constitutes an evaluation of the resistance of the

104-ohm standard i?#505.

5. Frequency Dependence of Bridge

Resistors

The measurements of the a-c-d-c resistance dif- ferences of Gx and G2 were made by comparing them

with a standard resistor of simple geometry having

predictable frequency characteristics, using both

direct current and 1592 c/s alternating current. Figure 18 shows the cross section of such a standard

- Table 2. Resistance of 10* ohm d-c resistor §505

Based on 1 pf reference capacitor #NBS 89790-B neglecting a-c-d-c differences

Date (1960) .R#505 Average phase angle of 10* ohm a-c resistors

May 5 May 10 May 11 May 19 May 20

10* ohms —73. 6 ppm

-73.4

-74.3

-74.1 -74.0

Microradians +15.0

+15.4

+14.8

+15.4 +15.2

Average —73.9 ppm +15.2

True value of 1 pf reference capacitor = l pf —54.0 ppm.

Resistance of i?#505=10' ohms —19.9 ppm.

Date (1960) i?#505 Average phase angle of 10* ohm a-c resistors

July 18 July 19 July 28 July 29

10< ohms —72. 2 ppm

-71.8 -72.5

-71.6

Microradians +14.7

+15.0

+14.8

+14.7

Average —72.0 ppm +14.8

True value of 1 pf reference capacitor=l pf —51.4 ppm.

Resistance of i?#505=10» ohms -20.6 ppm.

375
-' TO DETECTOR Figure 18. A-c-d-c resistance transfer standard.

whose resistance element consists of a single straight

wire 0.0008 in. in 'diameter and 9.5 in. long having a resistance of 1000 ohms. The structure is cylindrical with the resistance wire down the center. The large phase angle expected from a conventional

coaxial line resistor is eliminated by the use of a set of guard rings, g, about 4 in. in diameter, which sur- round the resistance wire. The potentials of the

guard rings are defined with a tapped inductor L

placed between the line terminal and ground. The

inductor taps are placed to provide a uniform potential gradient down the center of the cylinder in

the absence of the resistance wire. 4 When a bridge containing the transfer resistor is balanced with the resistance wire in place, the detector voltage is zero, and the potential gradient

along the wire is still uniform. Since the wire does

not change the electric field within the resistor

assembly, we may expect no capacitance contribution to the resistor phase angle and a-c-d-c resistance

difference. The effectiveness of the guard rings may

be checked by connecting adjacent guards in parallel to cut the effective number of them in half, and by

suitably reconnecting them to the tapped inductor.

If the apparent resistance and phase angle measured

in these two ways are the same, we may assume

that the number of guard rings is sufficient to eliminate the effects of capacitance within the resistor. In practice it is found that even with all of the guard

rings grounded, the measured a-c resistance is equal

to the a-c resistance with all guards connected

properly to the tapped inductor, and the phase

angle change between the two-guard arrangements

is less than 3 microradians.

The inductance of the standard resistor was

measured by replacing the fine resistance wire with

a larger low resistance wire and measuring, with a Maxwell-Wien bridge, the inductance including that

of the coaxial leads between the junction points

* Similar arrangements applied to high voltage transformer testing have been

described by Silsbee 110) and Wcller [11].

Figure 19. Circuit for comparing the transfer resistor with the bridge resistors, using ac.

of the bridge center used in the measurements to be reported below. A simple computation [12]

yields the difference between the inductances of the

fine wire and the low resistance wire. We find for the series inductance of the resistor attached to the

bridge center L=l.l microhenries. This inductance

has a negligible effect on the a-c resistance, but produces a phase angle of 11 microradians.

Figure 19 shows the bridge used for the a-c comparison of the 1000-ohm transfer resistor, RT, with

Gi OV-G2- The 10: 1 transformer is the one used for

the capacitance step-up, and has a known ratio. A

balance is observed on detector 1 with the resistors

disconnected at "X" to compare the 100-pf capacitor

with the 10-pf capacitor. When the resistors are placed in the circuit and both bridges are balanced, a change in the balance at detector 1 indicates a change in the transformer ratio between the junctions A and B, and allows the loaded voltage ratio

between these points to be calculated. This figure

combined with the readings of the admittances re- quired to balance detector 2 allows the a-c ratio of the resistances oi t G\ or Gi to the resistance of the

1000-ohm transfer standard to be determined, and

also allows the phase angles of G\ and 6?2 to be

determined from the known phase angle of the transfer standard.

Figure 20 shows the bridge used for comparing

the d-c resistance of the transfer standard with Gx or

6?2

. Since the tapped inductor of the transfer resistor presents a low d-c impedance to ground, it must be removed from the circuit for the d-c measurement. With this done, the leakage resistances

Ru R2 , and R3 are all high. Series resistances rx and r2 in connecting wires are small. Two balances

are made with points 1 and 2 grounded in turn. The direct reading ratio set readings are called Ai

and A2 , respectively.

The bridge is then connected as in figure 20,

where /?/ and 7?3

' are low resistance "shorts."

The bridge is balanced with the galvanometer con- nected to 3 and 4 in turn ; the readings of the DRRS

are called A3 and Ait respectively. An analysis

shows that for Gu p n _7?#400 ri , lOO(A-A) , \0(Ao-A<)-] HtGi -R¥505 1

1+ 115 + UB~~J

376
G, OrG 2 (lO.OOOftl

TRANSFER

RESISTOR UOOOfll 1

B

1 " LINK

r, (SMALL)

IO 4 ilST0 -WWiooo std Figure 20. Circuits for comparing the transfer resistor with the bridge resistor, using dc.

Provided that the resistance ratio of the d-c

standards is known, measurements of this form

yield the a-c-d-c differences of both Gx and G2

. Since

the 104-ohm resistor #505 was used for both this

measurement and for the direct d-c comparison

between it and Gx and G2 in connection with the fre- quency dependent bridge, one finds that resistor #505

serves only to maintain the unit between the two

measurements, and that in effect the 10:1 trans- former is used five times to step up the impedance of

the 1-pf standard capacitor to the 1000-ohm

transfer resistor. The d-c step-up is then needed

only from the bank of one-ohm reference resistors to the 103 -ohm d-c standard resistor #400. An error

in the d-c step-up from 103 ohms to 104 ohms would

cause no error in the final value assigned to the one- ohm reference resistors, but would only cause an

error in the measured a-c-d-c differences of 6X and G2 , which in this sense is not needed for our work. In fact, the only reason for including 104- ohm #505 in the measurements is that it is much

more stable than Gu G2 , and the transfer resistor. The use of 104-ohm #505 allows the complete determination to be broken up into smaller steps which

may be completed more quickly and with higher

accuracy. Subject to the above comments, we find

that Gi and G2 have identical a-c-d-c differences

and that their a-c resistances are 0.85 ppm higher

than their d-c resistances. The average of their

phase angles is found from the a-c measurement using

the phase angle calculated for the transfer resistor

to be 16.6 microradians (capacitive).

6. Conclusion

Combining the results of table 2 with the measured

a-c-d-c difference of Gx and G2 we find i?#505 = (104 ohms — 19.4 ppm). Comparison of this resistor

with the NBS bank of one-ohm reference standards

yields the value (104 ohms —21.7 ppm) in terms of

the ohm as adopted in 1948 [13] and as maintained

at NBS. Our measurements therefore indicate

that the NBS unit of resistance is 1.0000023 ohms.

Two values have been obtained for the phase angles

of the 104 -ohm woven-wire resistors; 16.6 microradians from the computed phase angle of the straight

wire a-c-d-c transfer resistor and 15.0 microradians

based on the assumption that the 1-pf reference

capacitor has no phase angle. The discrepancy of

1.6 microradians may be ascribed to the actual

phase angle of the 1-pf capacitor, an error of 0.16

microhenry in the measurement of the transfer

resistor series inductance, or a combination of these

two effects. The agreement is considered satis- factory in view of the uncertainties in these measurements.

Table 3 summarizes the steps of this determination and lists the estimated uncertainties associated

with each step. The estimates are in most cases

little more than guesses, since they must include

allowances for possible systematic errors. The

estimates are in all cases given as "50 percent

errors," meaning that in the experimenter's judgment the chances of the error being greater or less than this amount are approximately equal. It is judged to be almost certain that the error is less than

four times the stated 50-percent error. No allow- ance is made for the possibility of error in the as- signed value of the speed of light, which is taken as a predetermined constant, c =2.997925X 1010 cm/sec.

With the exception of the speed of light, all measurement uncertainties may be reduced with

suitable refinements in measurement techniques. It

is believed that with the development of an improved

cross capacitor and with minor changes in the rest of

the equipment, the measurement uncertainties not

including the contribution from the speed of light

may be reduced below 1 ppm.

Table 3. Uncertainties associated with steps followed in this determination

Measurement of the reference capacitor in terms of the computable

capacitor, neglecting the contribution from the uncertainty in the speed of light _ Instability of the reference capacitor Ten-to-one step-up using alternating current (times 5)._ Frequency dependent bridge — _ Dependence of bridge resistors upon frequency Step-down from 103 ohms to 1 ohm using direct current Measurement uncertainty neglecting the uncertainty in the speed

of light

56% error

ppm

2.0 0.3

.1.3

.3

.3

377
7. Comparision With Other Determinations

In order to compare several recent evaluations of

the 1948 resistance unit with each other, it is neces- sary to know the relative values of the units in terms

of which these evaluations were made. Figure 21

shows the results of comparisons made at the Inter- national Bureau of Weights and Measures (BIPM)

of resistors embodying the units as maintained by

NBS (tt EU), NRC (QCa ), and NPL (fl CB) in terms of the

unit as maintained at the International Bureau

(Gbipm) [14, 15, 16, 17]. No information is yet available concerning the 1960

intercomparisons, so it is impossible to compare the

present work with 0Bipm at this time. For this reason and for consistency in the table to follow, all measurements will be referred to the unit as maintained at NBS, flEU. Table 4 contains values assigned to ft EU by several

recent investigations [18, 19, 20]. The values have

been adjusted to allow for the difference between

S2EU and the units used for the measurements. The

results obtained by Thomas et al have been referred

to flfix; instead of the international ohm, and also revised upwards by 3 ppm. This revision represents

the results of a new evaluation of the current distri- bution in the primary winding of the mutual in- ductor, and is based on measurements of the dependence of resistivity upon strain in a sample of the

wire used for the winding. These measurements

were made in 1956 -by Wells [21]. The revised

current distribution correction was also calculated

by him, but the result was not published.

The National Bureau of Standards was fortunate

in receiving a visit in April and May of 1960 from

Mr. G. H. Rayner of the National Physical Laboratory in England. Mr. Rayner helped perform the

May measurements reported in this paper, and

contributed greatly to the early success of this

method. His contributions are gratefully acknowledged.

Others contributing substantially to the work

described in this paper were Mr. John C. Clark, Mr.

Lai H. Lee, Mr. John Q. Shields, and Mr. Thomas E.

Wells.

1 1

1 I n bipm'

7 NBS (flEU) V — o

0 NPL

0 0 .A

0

1 PPM

|

• •

• NRC

t

i i 1

z z

o o o°2 K Q CO O W 5

CE — O

0 <z Pi

2 . CD 1 2 in O ll) CO UJ LU U

""(CI

a. t1948 1950 1952

( 1950)

1954

(1953)

1956

(1955)

1958

(1957)

(REPORTED)

(MEASURED)

Figure 21. Resistance units maintained by NBS, NPL, and

NRC in terms of the unit maintained at the International Bureau of Weights and Measures (Hbipm).

Table 4. Values assigned to the resistance unit maintained at NBS (Vbu) by several recent investigations

(See text.)

i-Jut© 01 rncusurc* Value of the Keierenee

_Laboratory resistance unit

merits S2eo maintained

at NBS

1938 to 1949 Thomas. Peter- NBS 0.999997

son, Cooter, <fc Kotter

1951 Rayner NPL 1. 000004

1957 Romanowski & NRC .999996 Olson

1960 Present paper NBS 1.0000023

[3

[4

[5

[6

[7

[9;

[io:

[ii

[12

[13

[14

[is;

[16

[17

[is:

[19

[20;

[21

8. References

Thompson, A. M., and Lampard, D. G., A new theorem

in electrostatics and its application to calculable standards of capacitance, Nature, 177, 888 (1956).

Lampard, D. G., A new theorem in electrostatics with

applications to calculable standards of capacitance,

Proc. I.E.E.. Mono. No. 216M, January 1957. Thompson, A. M., The cylindrical cross capacitor as a calculable standard, Proc. I.E.E., Vol. 106, Pt. B, No. 27, May 1959. Lampard, D. G., and Cutkosky, Ft. D., Some results on

the cross capacitances per unit length of cylindrical 3-terminal capacitors with thin dielectric films on

their electrodes, Proc. I.E.E., Mono. No. 351 M,

January 1960. Thompson, A. M., The precise measurement of small

capacitances, I.R.E. Trans. Instr., vol. 1-7, Nos. 3 and 4, December 1958. McGregor, M. C., et al, New apparatus at the National Bureau of Standards for absolute capacitance measurement, ibid. Cutkosky, R. D., and Shields, J. Q., The precision

measurement of transformer ratios, Paper presented

at the 1960 Conference on Standards and Electronic Measurements at Boulder, Colorado, IRE Trans. Instr. 1-9, 243 (1060).

Johnson, G. J., and Thompson, A. M., Intermodulation

in bridge detector amplifiers, Proc. I.E.E., Mono No. 210M, November 1956. Thomas, J. L., Precision resistors and their measure- ment, NBS Circular 470, 1948.

Silsbee, F. B., A shielded resistor for voltage transformer

testing. Scientific Papers NBS No. 516, 1925. Weller, C. T., A 132-KV shielded potentiometer for determining the accuracy of potential transformers, A.I.E.E. 48, 1929. Grover, F. W., Inductance Calculations, Van Nostrand

Co., Inc., New York, N.Y., 1946. Comite" International des Poids et Mesures, Proces- Verbaux Deuxieme Serie, TXX, 1945-1946.

Comite" International des Poids et Mesures, Proces- Verbaux Deuxieme Serie, TXXIII, 1952. Comite" International des Poids et Mesures, Proces- Verbaux Deuxieme Serie, TXXIV, 1954. Comite" International des Poids et Mesures, Proces- Verbaux Deuxieme Serie, TXXV, 1956. Comite International des Poids et Mesures, Proces- Verbaux Deuxieme Serie, TXXVI, 1958. Thomas, J. L., et al, An absolute measurement of resist- ance bv the Wenner method, NBS Research Paper

2029, 1949. Rayner, G. H., An absolute measurement of resistance bv Albert Cambell's bridge method, I.E.E., Mono.

No. 95, 1954. Romanowski, M., and Olson, N., An absolute determination of resistance, Can. J. Phys., 35, 1957.

Wells, T. E., Measurement of the resistance-strain rela- tion and Poisson's ratio for copper wires, Proc. Instr. Soc. Am., 11, 1956. (Paper 65A3-97)

378
NATIONAL PHYSICAL LABORATORY

TEDDINGTON, ENGLAND

SYMPOSIUM ON PRECISION ELECTRICAL MEASUREMENTS

17th - 20th November 1954

PAPER NO. 14

Alloys for Precision Resistors

by

C. Peterson

(National Bureau of Standards, USA)

379
14. Alloys for Precision Resistors

by

Chester Peterson

National Bureau of Standards U.S.A.

SUMMARY

The National Bureau of Standards has always been vitally Interested In the properties

of metals and alloys suitable for use In precision resistors and has developed standards

known as the Rosa-Type and the Thomas-Type. This paper Is a condensation of a longer

unpublished manuscript which was Intended to serve as an historical summary and bibliographical guide for other workers In the field. During the decade 1930-40 Thomas did

much valuable work on resistance alloys which should be continued and expanded. One

object of such a continuing programme Is to produce stable standards, 10 ohms and higher,

having little temperature dependence in the range 20° to 300C. The present paper

summarizes the performance of 15 Thomas-type lohm resistors constructed in 1933; the

unit of resistance In the United States Is maintained by ten of these standards.

INTRODUCTION

When applied to resistors the word "precision 11 has a double connotation. In the

electronics field It signifies a resistor accurate to better than a few percent, whereas

In the electrical measurements field It means a resistor adjusted within 0.01% of its

nominal value, and of such quality and stability as to maintain this value over long

periods of time. In this paper we shall deal only with alloys suitable for use In the

latter field although a few of these alloys have also found application in the broader

electronics and communications fields.

Nearly a century has passed since Matthlessen conducted his classical researches on alloys suitable for the construction of resistance standards. During the Intervening

years numerous workers have investigated various phases of the problem and new alloys

adapted to particular uses have been developed. Intermittently during a period of over 25 years one of the most Indefatigable workers In this field was an Englishman, labouring In the United States, whose results were best publicized by the Germans. This was Edward Weston who developed for use in current shunts an alloy which later became known

as "manganic. This alloy deservedly enjoys much favour, although to this day it is not

fully understood what makes "good" manganln for precision standards.

This paper does not attempt to cover all aspects of the field. The prime objectives

here are to assemble a key bibliography, to present chronologically highlights In the

development of the art, and to point out gaps in our knowledge where further work might

advantageously be undertaken. This paper deals primarily with resistance alloys, although

some discussion of resistance standards Is inextricably Involved.

REQUIREMENTS FOR PRECISION RESISTORS

To be useful In precision resistors, alloys should possess the following properties:

(a) They should have a low temperature coefficient of resistance over as wide a temperature range as possible.

(b) They should, particularly for d. c. applications, have low thermoelectric power

against copper.

(c) They should be stable over long periods of time.

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380
(d) They should be workable by conventional wire and rolling mill practices.

(e) They should be easily soldered and welded.

In order to evaluate a given alloy with respect to the above properties It Is necessary to construct with it a resistor of some sort, be It a heavy four-terminal rod

or a coll of fine insulated wire wound on any one of a number of different coil-form

materials. Herein lies one of the difficulties In comparing the results of one Investigation with another, because the results obtained may depend to a notlclable degree upon the

form of the resistor and the distribution of stresses In the alloy. Frequently differences of opinion between producer and purchaser can be traced to these circumstances.

It Is Known, for example, that If two resistors are constructed using fine wire, one on a brass spool and the other on a steel spool, but otherwise Identical In construction, the

former may have a positive temperature coefficient, whereas the latter may have a negative

coefficient. The effect Is due to the differences In thermal expansion of brass, steel,

and the alloy In question. The point here simply Is that some uniformity of procedure

should be adopted whenever possible when comparing the resistance properties of various

alloys.

CHRONOLOGICAL DEVELOPMENT

The copper-nlclcel-zlnc alloy known as "German-silver" Is probably the oldest alloy

to have been used In the construction of resistance apparatus. It has a mean temperature

coefficient of resistance of about 450 parts per million per °C (ppm/°C) between the Ice and the steam point; It has considerable thermal e.m. f. against copper and Is not very

stable In Its resistance properties. It was the standard of comparison with respect to which early workers sought to find Improvements, and continued to be used until about the

turn of the century. It was eventually shown that the element zinc was objectionable In this alloy.

Literature published before the year 1900 frequently mentioned an alloy called

"Platinoid". This trade name was misleading because the material contained no platinum.

The alloy was essentially German-silver with small amounts of tungsten added.

The first worker to make a systematic Investigation of the suitability of various

alloys for use in resistance standards was A. Matthlessen who did much of his work under

the auspices of the British Association Committee on Standards of Electrical Resistance.

Matthlessen published a number of papers on the electrical properties of metals and alloys

during the decade 1855-65, of which two (refs.1,2) may be cited here. Much of

Matthlessen' s work on resistance alloys Is also published In appendices to reports of the

B.A. Committee (ref.3). The best alloy developed had a composition 2/3 silver and 1/3

platinum. This alloy had a temperature coefficient of about 325 ppm/°C and was the alloy

which the committee adopted for use in constructing the British Association type of

standard resistor. The work of Matthlessen can be summarized by quoting his concluding

remarks at the end of Appendix B In the Second B.A. Report (1863) as follows: "It will be

observed that I have not yet been able to find an alloy whose conducting power decreases

between 0° and 100°C less than that of the alloy of silver with 33.4%. platinum; and

from results obtained In this direction In conjunction with Dr. Vogt, I am of the opinion

that there will be great difficulty in doing so. We have already tested upwards of 100

alloys, and It is curious how few we have found whose conducting power varies less than

that of German-silver between 00 and 100°C.

In reptrospect one wonders why Matthlessen did not hit upon a combination which had

a temperature coefficient lower than 325 ppm/°C, but It should perhaps be borne In mind

that metals such as manganese and chromium, even In Impure form, were a rarity in his day.

Furthermore, Matthlessen seems to have laboured under the misapprehension (ref.4) that no practical binary alloy could have a temperature coefficient much smaller than the minimum

values he observed.

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To Edward Weston belongs the credit for producing the first resistance alloy having

substantially a zero temperature coefficient. At the time this was a remarkable achievement, particularly when It Is recalled that, In that day, metals by definition were materials which always had positive temperature coefficients of resistance. Unfortunately

for posterity, Weston was never Inclined to publish the results of his researches. It Is known that his Interest In resistance alloys covered a period of 25 or more years extending from about 1886 to at least 1910. His most Intense effort on alloys was made In the

years Immediately following the establishment of his private research laboratory In Newark,

New Jersey, In 1886. His Interest In alloys, arose out of some practical difficulties In connexion with the development of his portable Indicating Instruments. He needed a resistance material for voltmeter multipliers and ammeter shunts which did not vary In resistance with changes In temperature. Early shunts were made with German-silver.

A simple calculation shows that energy dissipation sufficient to produce a 40°C temperature rise will change the resistance of such a shunt by almost 2%, an amount clearly

Intolerable when the overall accuracy sought was a fraction of 1%. Weston and his assistants systematically compounded hundreds of alloys (ref.5) among which was found an alloy

of copper and nickel which actually had a negative coefficient. The firm of

Basse & Selve, of Altena, Germany, later coined the name "Konstantln" , and hence our name

"Constantan", for this low temperature coefficient alloy having the composition 60 copper40 nickel. This alloy was not entirely satisfactory because of Its large thermal e.m. f. against copper. Eventually Weston produced alloys containing manganese which proved

superior. Two brief news Items appearing In the technical press In 1888 (refs.6,1) described these as zero temperature coefficient alloys having the compositions 70 copper30 ferromanganese, or 65 copper-25 to 30 ferromanganese - 2.5 nickel. No further details

were given.

The word "Manganln" was coined by the Germans to describe a copper-nickel-manganese

alloy of rather specific composition. The Physlkallsch-Technlschen Relchsanstalt

(P. T. R. ) was founded In 1887 and within a year after Its establishment a definite programme

for the development of resistance standards was under way. The precise relation between

Weston's work and the Investigations of the P. T. R. Is difficult to unravel at this late

date, but In 1888 Weston's name was well known In electrical Industry throughout the world

and German technicians were aware of his researches on resistance alloys. In his papers

on resistance alloys published In 1889 and 1892 Feussner (refs.8,g) makes brief mention

of Weston's alloys. In the latter paper Feussner states that an electrical manufacturer

In Berlin attempted to make an alloy In accordance with Weston's patent specifications

using ferromanganese, but the product was brittle and could not be drawn to wire.

Feussner was of the opinion that Iron wa^ responsible for the brlttleness.

The Isabellen Metal Works* In Dlllenberg, Germany, had for some time used a coppermanganese alloy for certain machine parts, Feussner ordered from this company samples

of that alloy with and without additions of nickel. In his 1889 paper Feussner gave

preliminary values for the electrical properties of these alloys. For nominal compositions 70 copper-30 manganese, and 73 copper-24 manganese-3 nickel, Feussner reported

temperature coefficients of +40 and -30 ppm/°C respectively. The chemical composition

of these alloys was not determined by analysis; consequently the amount of Iron and other

Impurities was not known. Historically speaking, the situation Is beclouded by the fact

that Weston was reputedly dissatisfied with the Inability at that time of American wire

mills to draw his copper-nickel alloy ("constantan") to the fine gauges he wanted.

According to Woodbury. [loc. clt. p. 168] Weston employed German mills for this purpose

and the Isabellen works may have been one of those so employed. This company coined

the name "Westonln" for the copper-nlckel-manganese alloy made to P.T.R. specifications,

but later the more descriptive name "Manganln" was adopted. Whatever the precedence of

development may have been, no less a person than von Helmholtz stated In 1892

[loc. clt. p. l?9] that "the discovery of a metal whose resistance diminishes with

temperature was made by an American engineer" (Weston).

* This firm Is still In business under the name Isabellen-Hutte Heusler K.-G. , (16) Dlllenburg

(Hessen) Germany.

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The remark, by von Helmholtz quoted above was made at the nineteenth meeting of the B.A. Committee In Edinburgh, 1892, He and other representatives from Germany, France,

and the United States were present by Invitation to consider the problem of establishing

Identical electrical standards In various countries. Llndeck presented a paper on wire

standards of resistance describing some of his work at the P.T.R. This paper appeared

In English as an appendix (ref.io) to the report covering that meeting.

In 1895 the P.T.R. published In Its own Journal a paper (ref.n) summarizing all the

work done on wire standards since the beginning of the project In 1888. That paper was

a landmark In the history of resistance alloys and standards, and merits some consideration

at this point. The paper consisted of four major parts, of which only the first and the

last are of Interest here. The first part discussed the alloys Investigated and the last

part gave data on the stability of standards constructed at the Relchsanstalt. The alloys

Investigated were divided Into two groups: (a) the commercially available alloys platlnumlrldlum, platlnum-s liver, german-s liver, nlckelln, and patentnlckel; (b) new alloys

Identified by the names "manganln" and "constantan". Platinum- Iridium was not actively

considered because of Its cost and high temperature coefficient; the authors had a poor

opinion of Matthlessen' s platlnum-s liver alloy because the samples they had were inhomogeneous and brittle. Considerable attention was paid to the last three alloys In group

(a), particularly "patentnlckel". This last-named material was a coinage alloy

(essentially 75 copper-25 nickel) used by the German mint, and with It a temperature

coefficient of resistance as low as +170 ppm/°C was obtained. The authors recognized

that the principal limitation on the accurate measurement of resistance was the temperature coefficient of the resistance alloy used. They observed that alloys of copper and

nickel without zinc had stable properties and that lower temperature coefficients were obtained with Increasing amounts of nickel. Accordingly they directed their further

researches to binary copper-base alloys containing various amounts of nickel or related

elements. They were aware of the fact that Weston had obtained negative coefficients

with alloys Involving copper and manganese. For their Investigation on new alloys they

obtained a series of copper-manganese alloys from the Isabellen works, and a series of

copper-nickel alloys from the firm Basse and Selve. In the first alloy series they

found that a manganese content of seven or more percent gave low temperature coefficients

and a small negative thermal e.m. f. against copper. in the second series they found

that negligible coefficients were obtained with nickel contents between 40 and 60%; the

thermal e.m. f. of these alloys was about 40 (iV/oc with respect to copper. The authors

therefore probably reasoned that by adding a small amount of nickel to the coppermanganese alloy a material might be had which would possess a low temperature coefficient

of resistance and negligible thermal e.m. f. against copper. The proportions finally

adopted were approximately 84 copper-12 rnanganese-4 nickel. The alloy of this rather

specific composition made by the Isabellen works for the P.T.R. is the alloy which became

known as "manganln".

The last part of Feussner and Llndeck' s paper gave observations on the stability of

45 standard resistors having decimal values ranging from 0.1 ohm to 10 000 ohms. These

were constructed according to the authors' design and became known as the "Relchsanstalt

Type" of standard resistor. In tbls group of resistors 25 changed less than 0.01%

during a period of l£ to 3 years. The stability measurements referred ultimately to mercury standards on which the P.T.R. concurrently did much work.

The P.T.R. actively pursued Its work on manganln resistors and set up working standards covering the range 0.001 to 10 000 ohms. Most of these standards were manufactured

to P.T.R. specifications by Otto Wolff* who limited his efforts to the production of

standard resistors and resistance apparatus only, and achieved In this field an International reputation for the quality of his workmanship. Reports on the stability of these

standards are contained in papers by Jaeger and Llndeck (ref.12).

* Otto Wolff senior died of natural causes during World War II. The business of the firm Is carried on by a son, also named Otto Wolff. The firm Is located In the Western Sector of Berlin and their present address Is: Abbe-Strasse 12, Berlln-Charlottenburg 2, Germany.

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As a result of the thoroughness of the P. T. R. investigations and the excellence of

Wolffs manufacture, manganln standard resistors of the Relchsanstalt type became firmly

established in scientific and testing laboratories throughout the world. When the

National Bureau of Standards was organized In 1901 it acquired Wolff resistors with

Relchsanstalt certification as rapidly as funds permitted. During the period 1907-

1909 (ref.13) tne unlt of resistance In the United States was maintained with a group of

ten lohm and seven o.lohm. standard resistors of this type. After May, 1909, the unit

was preserved with a sealed type of resistor developed by Rosa. Twice In the history

of the N. B. S. Improved resistance standards have been developed by changes In design and

fabrication procedures rather than by changes in the resistance alloy employed. The

first improvement was the development of the N. B. S. type of sealed resistor alluded to above, and described in a paper by Rosa (ref.14) ln 1908. The second improvement was

the development of the double-wall type of resistor by Thomas around 1930, regarding which more will be said in a subsequent portion of this paper. In each case the alloy used

was manganln.

Before World War I, Instrument manufacturers ln the United States Imported practically all their constantan and manganln from Germany. When this source of supply was

cut off, the stocks on hand were eventually depleted and domestic sources of supply had

to be developed. The essential difficulty was not inability of domestic producers to make sound wire; the problem was more one of specifications. The foreign product

varied considerably ln Its properties and users differed ln their opinion regarding

desired requirements. This situation stimulated a number of Investigations among which

may be mentioned the work, of Hunter and Bacon at Rensselaer Polytechnic Institute

(ref.15) 311(1 F' E« Dasn at tne Leeds and Northrup Company (refs.16, ij). The results of

these investigations showed that the peak of the resistance- temperature curve for manganln

could be placed near 25°C If the total iron content of the alloy was raised somewhat

above that which resulted from the unavoidable Iron content of the impure manganese

employed. However, ln comparing the results of Hunter's work with thoseof Bash, there seems to be some conflict of opinion regarding the effect of Iron. Hunter states that

the samples to which no extra iron was added had maximum resistance around 40°C, and that

with increasing additions of iron the maximum resistance is reached at lower temperatures.

Bash states that with small amounts of iron the maximum of the resistance- temperature

curve occurs at low temperature, and that increasing amounts of Iron up to 1.5% raise the

location of the peak; Iron beyond 1.5% apparently brought the peak down again. The

situation may be affected by the amount of silicon or other impurity present, but ln any

event both authors agreed that a total of 1.0 to 1.5% iron placed the peak of the curve near 25°C. Most manganln manufactured domestically after 1920 therefore contained extra

Iron, particularly If intended for use ln precision resistors as distinguished from

current shunts.

It should be noted that the manganese used before about 1937 usually contained

several percent of Impurities; this might account for some of the discordant results

reported by Hunter and Bash.

In 1925 Pilling published a paper (ref.18) presenting the results of a more or less

cursory examination of some electrical properties of the copper-nickel-manganese series

of alloys, tracing the major effects of composition on resistivity, temperature coefficient and thermoelectric power. Pilling covered a large portion of the ternary diagram

and verified that alloys with zero or negative temperature coefficient could be obtained over a broad range of compositions, Involving about half the area of the diagram. However, when consideration was limited to workable alloys having low thermal e.m. f. against

copper, the alloys of Interest centered. around two fields: (1) a group containing less

than 10% of nickel, with from 10 to 35% of manganese; (2) a group containing about 50%

of nickel and 30% of manganese. The first of these groups Includes the manganln- type

alloys; trie seccnd was marginal ln workability and r.as apparently never teen studied ln detail.

No survey of the present type would be complete without reference to the Investigations of J. 0. Llnde made in Sweden at the Tekniska Hogskola, Stockholm, during the period

1928-38. Llnde was Interested ln the electrical properties of the binary solid solutions

formed when pure gold, silver or copper were diluted with small amounts of other elements.

His results were expressed as the proportional Increase ln resistance per atom percent of

diluent, and the data were arranged to show how these numerical values depended upon the

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relative location of the solvent and diluent metals in the periodic table of the elements.

The effect of temperature was included in the investigation. Most of Llnde's work is published in Annalen der Physik, to which one reference may be made here (ref.ig). In 1S39 all of Llnde's work, in this field was assembled and published (in German) in a paperbound volume (ref.20). Although Linde Included only a few percent of diluent metal in

most cases, his work was of great value because it provided the stimulus for later practical development work on gold-chromium alloys in the N.B.S. by Thomas, and on silvermanganese alloys by Schulze at the P.T.R.

In 1928, Thomas began an investigation at the N.B.S. aimed at determining the cause

of erratic variations in the values of the standards used to maintain the unit of resistance. Since May, 1909, the unit in the United States had been preserved with a group of

iofcm sealed resistors of the Rosa type. Although these resistors were of high quality,

experience with them during the ensuing decades revealed that individual resistors might

change several ppm/ year with respect to the average of a group of ten. Manganln colls

for these standards were Insulated with silk and were prepared and baked according to the methods used at the P.T.R. and described In the papers of Feussner and Llndeck of that

organization. Because of the silk Insulation the baking temperature was limited to 140°C. This baking procedure was necessary In order to dry the shellac and to help

stabilize the resistance; no significant annealing of the wire took place, although some partial stress relief undoubtedly occurred. From Brldgeman's work (ref.21) it is known

that resistivity is influenced by stress. Thomas reasoned that If he annealed the wire

at a temperature sufficiently high to remove all stress better stability would be obtained.

Accordingly he used bare manganln and annealed it in vacuum at 550°C (ref.22), supporting

the coll on a steel mandrel of the same diameter as the spool on which the finished coll

was. to be mounted. The annealed coll was anchored to a silk-insulated tube with linen

thread, after which it was coated with shellac and baked for an hour at about 800C. The

finished coll was sealed In a double-walled container. These resistors gave very encouraging results; Thomas then modified the design slightly by increasing the size of the wire

and the container. A group of these large double-walled resistors was made at the N.B.S.

In 1933 and a review of their performance was published 13 years later, (ref.23). These

resistors proved to be superior to any previously developed; consequently the unit of

resistance in the United States was preserved with this type of standard exclusively,

beginning In 1939. With respect to the mean of ten the value of any individual standard

changed only a few parts In ten million per year. To carry forward the performance data

given in Tables 1 and 2 of ref.23 the observations made since 1945 have been assembled

in Table 1 below. This table has been prepared on the same basis as that employed by

Thomas In the reference cited, i.e. it Is assumed that the average value of the 10

resistors maintaining the unit had remained constant. As noted, some resistors have met

with accidents, yet the average change of all 15 without regard to sign was only 1.5 ppm

over a period of 21 years.

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TABLE I

Stability of lohm double-walled manganin resistors.

change In units on January 1, 1948, Is omitted.)

(In this table the effect of

Standard

No.

Change In resistance, In microhms, from 1933

1S33 1945 1946 1947 1948 1949 1950 1951 1952 1954

68 0 *0.1 -0.1 -0.4 -0.4 -0.6 -0.6 +3. 7 (a) +3.6 +3.7

70 0 +0.3 +0.5 +0.4 +1.0 +0.9 +0.8 +0.9 +1.0 +1.1

74 0 -0.7 -0.3 -0.6 -0.8 -0.7 -0.8 -1.1 -1.2 -1.4

77 0 -1.0 -1.1 -1.1 -1.1 -1.1 -1.3 -1.3 -1.4 -1.3

78 0 +1.0 +1.1 +1.3 +1.5 +1.5 +1.6 +1.7 +1.8 +1.8

81 0 -0.9 -0.6 -0.6 -0.5 -0.7 -0.7 -0.7 -0.7 -0.9

82 0 -0.4 -0.0 +0.2 +0.1 +0.4 +0.2 +0.1 +0.2 +0.1

84 0 +0.6 +0.6 +0.3 +0.6 +0.4 +0.6 +0.4 +0.6 +0.5

89 0 +0.1 -0.2 -0.1 -0.6 -0.2 -0.4 -0.3 -0.3 -0.5

90 0 +0.5 +0. 7 +1.1 +0.8 +0.8 +1.0 +1.1 +0.9 +1.2

67 0 -1.5 -1.4 -1.0 -0.8 -1.0 -1.5 -1.3 -1.6 -1.7

69 0 +1.9 +4.7 (b) +4.6 +4.5 +4.4 +4.4 +4.3 +4.3

72 0 +0.4 +0.4 +2.8 +2.7 +2.5 +2.6

73 0 -0.9 -1.2 -1.5 -2.0 o.o(c)

83 -2.1 -1.5 -1.5 -1.9 -1.3

Notes: (a) This resistor was accidentally dropped In January 1951, which caused a sudden

Increase in resistance of about 4 microhms; Its value has remained constant since that date.

(b) (c) These resistors were employed In International lntercomparlsons Immediately

preceding the date noted. In 1953 Nos.72, 73, and 83 made a round trip to France by air via diplomatic mall pouch; In each direction of travel they

Journeyed In 3 separate flights and were not given preferential treatment.

The resistance of No. 73 Increased 1.3 microhms during Its absence from

Washington.

The manganin employed In these large double-walled resistors has been of much

Interest and the source of numerous Inquiries. The wire was obtained from the

Driver-Harris Company. In 1933 and was manufactured before the advent of electrolytic

manganese. It was Known that the 550°C anneal would raise the temperature coefficient;

the producer was therefore requested to supply, If possible, manganin wire which had a slightly negative coefficient. Two lots of wire were supplied; the wire appeared to be quite soft, and as received had a zero temperature coefficient at about 20°C. The

finished standards had small coefficients, varying from zero to +3ppm/°C at 25°C. The

chemical analysis of this wire Is given In table 2 overleaf. Samples A and B were from

lots 1 and 2 respectively, and were analyzed at the N.B. S. In 1933. Sample C came from

the remains of lot 1 or 2 used by Thomas In 1933, and was given to the Leeds and Northrup

Company In 1947, which organization redetermined the composition. Their results are Included to show silicon content.

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TABLE 2

Analyses* of Manganin Used in ff.B.S. Double-Called Standards

Sample A 3 C

Copper

Manganese

82.2

13.2

4.2

0.3

82.7

12.1

4.6

0.5

82.38

12.98

4.22

0.246

0.042

Nickel

Iron

Silicon

* All values are In percent by weight

During the decade 1930-40 Thomas Investigated the resistance properties of the following alloys: gold- chromium (ref.24), gold-cobalt (ref.25), and copper-manganese-alumlnlum

(ref.26). A brief summary of the resistance properties of these alloys will be given

here, but the Interested reader should refer to the original papers for further details.

The Investigation of the gold alloys was suggested by Llnde's work, referred to above.

Thomas found that the resistivity of a gold-chromium alloy containing 2,1% chromium by

weight could be made Independent of temperature to less than 1 part In a million In the

range 20° to 30°C. This was achieved by a suitable baking procedure. Most alloys

exhibit an Increase In temperature coefficient at 25°C and a decrease In resistivity when

baked at elevated temperatures. Gold-chromium behaves In an unusual fashion Inasmuch as baking produces the opposite effect In that the resistance Increases and the temperature

coefficient decreases on baking. For alloys of the proportions mentioned It was found

that a linear relationship existed between the Increase In resistance and the decrease In coefficient when baked at 150°C. A 1% Increase In resistance was accompanied by a decrease of 26 ppm/°C In the temperature coefficient. By employing this relationship,

the coefficient could be adjusted to zero with comparatively few determinations of its

value because the resistance changes were large and could be measured and followed with

ease. The general shape of the resistance- temperature curves of these alloys Is well

Illustrated In figure 3 of a later paper by Godfrey (ref.27). The thermal e.m. f. of

gold-chromium against copper Is 7 or 8 u-V/oc at 25°C as compared with 2 or 3 uV/°C for

manganin against copper. Gold-chromium alloy shows to best advantage when used for

constructing standards of 10 ohms or more. Two lohm standards have been made using

heavy wire mounted In the large-type double-walled containers. Over a period of 4 years

the resistance of one has decreased 6.6 ppm while the other has Increased 1.6 ppm; the

stability of these Is apparently no better than the best manganin double-walled resistors,

and may In fact be not quite as good. The advantage of gold-chromium alloy lies In the

fact that the resistance of colls made with the small wire sizes appropriate to nominal

values In the range 10 to 100 ohms can be manipulated by baking, so that the temperature

coefficient can be made substantially zero over the Interval 20° to 30OC.

Llnde's work Indicated that 2% by weight of cobalt In gold would give an alloy having

zero temperature coefficient around 250C. Thomas therefore prepared a series of goldcobalt alloys having from 0.75 to 5% cobalt. He found that 10-ohm colls when baked for

18 hours at 140°C would have zero temperature coefficients If the cobalt content was about

2.2% by weight. The thermo-electric power of this alloy against copper Is very high,

being about 45 u.V/°C. The stability of a few resistors made with this alloy indicated

that the general performance was inferior to gold-chromium.

Alloys similar to manganin, but with the nickel replaced with about the same weight

percent of aluminium, first came to the attention of the National Bureau of Standards

around 1910. At that time Wilbur B. Driver donated samples of such an alloy which he

had developed as an electrical-resistance material. This material had a very low thermal

e.m. f. against copper and was of the type which later became known as "Therlo". The

composition of Driver's material was approximately 84 copper- 12 manganese-4 aluminium, with

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traces of Iron and nickel. A number of resistance standards were made from these samples

and kept under observation and In regular use for more than 20 years. The alloy used In these standards was not physically sound Inasmuch as Inclusions and Imperfections were readily visible. The performance of these standards was sufficiently good, however, to warrant a more complete Investigation of the resistance properties of alloys of this type.

Sucn an Investigation was finally made by Thomas In the early 1930' s. Thomas made a series of alloys containing from 80 to 95% copper with varying proportions of manganese

and aluminium, and he worked out a procedure for making sound material in small quantities.

The resistivity and temperature coefficient of these alloys were quite appreciably

Influenced by heat treatment. Baking at 140°C for 18 hours changed the resistivity by

over ic% In some cases. A number of compositions were found which gave substantially

zero temperature coefficient after baking. Of these alloys, that alloy which gave the

lowest thermoelectric power against copper was found to be 85 copper-9.5 manganese5.5 aluminium. Compared to manganln this alloy had approximately the same resistivity;

Its temperature coefficient could be made zero at 25°C; the change of the resistance with

temperature was about half tnat of manganln; and the thermoelectric power against copper

was only 10% that of manganln. The structure of this alloy was evidently complex because

the wire would become harder If It were heated at too high a temperature in the attempt to anneal It during the swaging process. Ductile cold-worked wire, after mounting on the

Insulated spool for the finished coll, would become brittle after the 140°C baking procedure. These effects may have been caused by a precipitation hardening phenomenon;

some metallographlc work should be Included In any future work on alloys of this type.

During the period 1933-41 Alfred Schulze In Germany did considerable work on alloys

for standard resistors at the P. T.R. ; seme of his work paralleled and confirmed that of Thomas,

particularly on the high- temperature treatment of manganln (ref.28) and on gold-chromium

alloys (ref.2g). Of more Interest here, however, are two other papers of Schuize's, the

first of which (ref.30) Is a summary of the properties of certain resistance materials

suitable for use In constructing standard resistors. That paper Is divided into three

parts: (l) resistance materials based on copper-manganese; (2) gold-chromium;

(3) peculiarities In physical behaviour of resistance materials. In part I the author

reviews the properties of manganln as a function of annealing temperature, and he Includes

data on three alumlnlum-bearlng copper-manganese alloys. Two of these were commercial

German alloys known as " Isabellin" and "Novokonstant" , while the third, called

"A- Leglerung" , was Intended to duplicate the composition adopted by Thomas In his paper

(ref.26) on copper-manganese-alumlnlumalloys. In agreement with Thomas, Schulze found

that these alumlnlum-bearlng alloys had resistance properties which were quite sensitive

to heat-treatment. He found that the alloy " Isabellin" (84 copper-13 manganese-3

aluminium) had a temperature coefficient of about -38 ppm/°C In the cold-worked condition,

but that after heating for 75 to 85 hours In a neutral atmosphere at 500°C the coefficient

could be made very small. Small coefficients could also be obtained with the other two

alloys by following a similar baking process. Schulze made experimental resistors from

all three alloys and obtained stable colls from " Isabellln" and "Novokonstant" In most

cases. However, with Thomas' alloy (85 copper-9.5 manganese-5.5 aluminium) Schulze was unable to produce stable resistors. This result Is not in agreement with results obtained

at N.B.S. and Is further evidence that alloys of this type possess some metallurgical

peculiarities which are not fully understood. Alloys of this type ("therlo") have many

desirable features and It appears that further study Is warranted.

In part II of the paper under discussion (ref.30), the author makes brief reference

to gold-chromium resistors containing 2.05% chromium by weight. These were baked at

about 2000C for 16 to 20 hours to reduce temperature coefficients to low values. Their

behaviour was entirely In accord with the observations of Thomas and Godfrey In the United States, In part III the author discusses the effect of atmospheric humidity on the resistance of

colls attached to supporting spools with shellac and describes a new form of resistor

employing a sealed container, a type of construction not heretofore used extensively In Germany. The early work of Rosa at the N.B.S. demonstrated the necessity for sealing

precision resistors against the effect of atmospheric humidity, and In this country

precision resistors of lohm and higher have been enclosed In sealed containers for the

(19649)

388
past 40 years. Indeed, experience In the United States has shown that the beneficial

effect of the high temperature anneal of manganln Is largely lost if the resistors are left open and exposed to the atmosphere.

The second of Schulze's papers to which particular reference will be made here deals

with the use of silver-base resistance alloys (ref.31). That paper Is of great Interest

because Schulze demonstrated that certain silver alloys could be made which had very small

temperature coefficients around room temperature, and which had other desirable properties

making them useful In the construction of precision resistance standards. From Table 8 In Llnde's book, (ref.20) It Is apparent that silver alloyed with 7 to 17 atomic % of manganese would have substantially a zero temperature coefficient. A few resistors using

alloys of this type have been made at the N. 3. S. during recent years; no systematic

study has as yet been made here, but the results support the conclusions of Schulze and

Indicate that further Investigation of silver-manganese alloys would be profitable. It

Is Interesting to note that although Matthlessen never succeeded In producing an alloy

having lower temperature coefficient than about +325 ppm/°C obtained with his sllverplatlnum alloy, Schulze was able to produce silver alloys having the largest negative

coefficients ever observed In metallic solid solutions as far as the author knows.

Schulze studied five alloys consisting of silver alloyed with from about 9 to 22% of

manganese and tin In various proportions. The alloy containing least silver had a com- position 78 silver - 13 manganese - 9 tin, and although Its electrical properties were very good, the alloy had such poor mechanical properties as to render It unsatisfactory

for extensive use. Three of the five alloys were found to be particularly useful In the construction of standard resistors. Their properties are summarized In Table 3 below.

TABLE 3

Electrical Properties of Three Silver Alloys

(From data by Schulze)

at i Nominal Resistivity, mlc rohm-cm „ „ Finished Re sistors , AUoy Composition cTwT H.T." a

' C * W' J

de/dT

A 8.78 Mn, bal Ag 32 28 -40x10-6 -0.85X10- 6 -0.04X10- 6 +2.5

B 10 Mn 8 Sn 82 Ag 55 51 -36X10- 6 +1.70X10-6 +0.23x10-6 +0.5

C 17 Mn 3 Sn 80 Ag 58 46 -105x10-6 +0.49X10- 6 +0.05X10- 6 +2

Notes: Data taken from ref.31. C.W. = cold worked. H. T. = heat treated. a and 3 are constants In the quadratic resistance- temperature equation. de/dT - thermoelectric

power against copper, In UV/°C, In the vicinity of 25°C.

The properties of these three alloys depended greatly upon their mechanical and thermal

treatment. The very large negative temperature coefficient of the cold-worked wire

could be changed to substantially zero by heating it In accordance with a schedule which

varied with the composition. Alloys A, B, and C were given the following respective

heat treatment: 12 hours at 2500c, 10 hours at 175° C and 18 hours at 270°C. The finished

resistors referred to were open colls on porcelain spools, of 1-ohm nominal value, using

wire of 0.5 and 0.6 mm diameter. No data are available regarding the long-time stability

of these resistors, but data taken over a period of 1 to 10 months indicated that they

remained constant within a few ppm. Schulze's work on silver alloys may be summarized

by stating that alloys of this type can be prepared which are responsive to heat treatment, and which can be processed In such a manner as to make their temperature coefficients

extremely small. The resistance-temperature curve Is much flatter than In the case of

(19649)

389
manganln; In this regard the alloy compares favourably with gold-chromium. The thermoelectric power against copper Is not greater than that of manganln.

In recent years two new resistance alloys of a modified nlckel-chromlum type have

been Introduced In the United States. They are produced by the Wilbur B. Driver Company

and the Driver-Harris Company under the respective trade names "Evanohm" (refs. 32, 33 J and

"Karma". They contain approximately 74 nickel and 20 chromium, the balance being alunln.lum plus Iron or copper. The thermoelectric power of these alloys against copper Is about

the same as that of manganln, but their resistivity is higher by a factor of 2.5 to 3 times. By a suitable annealing and baking process the 25°C temperature coefficient can

be made to fall in the range +20 ppm/oc. The resistance-temperature curve of these

materials Is fairly flat over a wide temperature Interval.

Mention should also be made of a new alloy sold by the C 0. Jelllff Mfg. Corp.,

known as "Jelllff Alloy 800". This alloy has a resistivity of approximately 800 ohms

(circular mil, ft), and Is composed of substantial amounts of nickel, chromium and

manganese, together with smaller amounts of molybdenum; It can be made to have a temperature coefficient of +20 ppm/°C in the vicinity of room temperature. The thermoelectric

power against copper Is about twice that of manganln. This material, like the modified

nlckel-chromlum alloys mentioned above, has greater strength than the copper-base alloys

and consequently fine wire can be handled with relative ease In automatic winding machines

employed for manufacturing wire-wound resistors used in considerable quantities In the

communications and seml-preclslon Instrument fields. The three new alloys mentioned are frequently used in constructing high-valued wire resistors. A limited number of precision standard resistors have been made In the N.B.S. using these new alloys. They have

satisfactory temperature coefficients but no great amount of data on long-time stability

has been accumulated. Some difficulty due to hysteresis effects have been experienced

with the alloys containing aluminium. These alloys are of sufficient interest to warrant

further study.

RECAPITULATION

In this paper reference has been made to the principal resistance alloys that have

at one time or another been used In the construction of precision resistors. It Is a tribute to the excellent qualities of manganln when one considers that this alloy still

finds wide favour after 60 years of use, although to some extent It Is being displaced by

the new high-res is tlvlty alloys of the modified nlckel-chromlum type. Many studies of

manganln-type alloys have been made, yet there is no adequate explanation for the instability that has occasionally been observed in some lots of this material. The effect

of residual iron and silicon in this alloy Is not fully understood. Unpublished results

of studies by Thomas on alloys of the. manganln type made with electrolytic manganese

Indicate that the exact proportions of copper, nickel, and manganese are not critical. A series of alloys of this type can be made with copper in the range 70 to 85% and all with

substantially zero temperature coefficient. In this series of alloys the thermoelectric

power against copper and the curvature of the resistance-temperature graph have values

depending on the composition; Thomas was unable to adjust the composition so as to simultaneously obtain negligible values for these properties. An investigation currently

under way in the N.B.S. alms to complete this work.

The work of Schulze on sliver alloys indicates that further investigation of modified

silver-manganese alloys should be made. Alloys with very little temperature dependence,

comparable to gold-chromium, appear to be attainable.

Alloys of the therlo type have long been of Interest because of their low thermoelectric power and because their resistance-temperature curve has less curvature than that

of manganln. More work needs to be done on these alloys to determine why they sometimes

become brittle and show Instability.

(19649)

390
A few standard resistors have been made In the N.B.S. using the hlgh-reslstlvlty

alloys Evanohm, Karma, and Jelllff Alloy 800. Further studies are required In order to determine their suitability for the needs of national standardizing laboratories.

References

1. A. MATTHIESSEN, On the electric conducting power of alloys, Trans. Roy. Soc. (London) I860, 150, 161.

2. A. MATTHIESSEN and C. VOGT, On the Influence of temperature on the electric

conducting power of alloys, Trans. Roy. Soc. (London) 151, 167 (1864). 1864, 151, 167.

3. British Association Reports on Electrical Standards. (Reprinted 1913 by Cambridge

Univ. press.

)

4. First B.A. Report (1862). appendix A. (See ref.3).

5. DAVID 0. WOODBURY, A measure for greatness, p. 168. McGraw-Hill, 1949.

6. Weston's new alloys for electric conductors. Elec. World, 1888, 12, 39.

7. Alloys for electrical resistances with no temperature coefficient, Science 1888, 12,

156. See also U.S. patents 381304 and 381305.

8. FEUSSNER and LINDECK, Metalleglerungen fur elektrlsche wlderstande, Zeits. fur

Instrumentenkunde , 1889, 9, 233.

9. FEUSSNER, Neue materlallen fur elekxrlsche messwlderstande, Elektrotechnische Zeits.

1892, 13, 99.

10. STEPHAN LINDECK, On wire standards of electrical resistance, 19th B.A. Report (1892),

appendix IV. (See ref. 3).

11. FEUSSNER and LINDECK, Die elektrlschen normal-draht-wlderstande der PhyslkallschTechnlschen Relchsanstalt, Wiss, Abh. der P.T.R. 1895, 2, 503.

12. JAEGER and LINDECK, Ueber die konstanz von normal-wide rstanden aus manganln, Zeits.

fur Instrumentenkunde, 1898, 18, 97; 1906, 26, 15.

13. F. B. SILSBEE, Establishment and maintenance of the electrical units, N.B.S. Circular

C475 (1949). See p. 6.

14. E. B. ROSA, A new form of standard resistance, Bui. BS, 1908-9, 5, 413, S107.

15. M. A. HUNTER and J. W. BACON, Manganln, Trans. Am. Electrochem. Soc, 1919, 36, 323.

18. F. E. BASH, Manufacture and electrical properties of constantan, Am. Inst. Mining and

Met. Engrs., 1921, 61, 239.

17. F. E. BASH, Manufacture and electrical properties of manganln, Am. Inst. Mmmgand Met. Engrs. , 1921, 61, 269.

18. NORMAN B. PILLING, Scir.e electrical prcjertles of copper-nickel-manganese alloys,

Trans. Am. Electrochem. Soc. , 1925, 18, 171.

19. J. 0. LINDE, Elekxrlsche elgenschaften verdunnter mlschkrlstalleglerungen, Ann.

d. Physik, 1932, 107, 219.

(19649)

391
20. J. 0. LINDE, EleJctrlsche wlderstandselgenschaf ten der verdunnten legierungen des

kupfers, sllbers und goldes. Gleerupska Univ.-Bokhandeln, Lund, Sweden, (1939).

21. P. W. BRIDGEMAN, The electrical resistance of metals under pressure, Proc. Am. Acad. Arts & Sc\. , 1917, 52, 537.

22. J. L. THOMAS, A new design of resistance standard, BS J. Research, 1930, 5, 295,

RP 201.

23. J. .L. THOMAS, Stability of double-walled manganln resistors, BS J. Research, 1946,

36, 107, RP 1692.

24. J. L. THOMAS, Gold-chromium resistance alloys, BS J. Research, 1934, 13, 681, RP 737.

25. J. L. THOMAS, Gold-cobalt resistance alloys, BS J. Research, 1935, 14, 589, RP 789.

26. J. L. THOMAS, Electrlcal--res lstance alloys of copper, manganese, and aluminium,

BS J. Research, 1936, 1 6, 149, RP 863.

27. T. B. GODFREY, Further data on gold-chromium resistance wire, BS J. Research,

1939, 22, 565, RP 1206.

28. ALFRED SCHULZE, Elektrlsche und thermlsche untersuchungen an manganln, Physik.

Zeits. 1937, 38, 598, 39, 300 (1938).

29. ALFRED SCHULZE, Uber die gold-chrom-wlderstandsleglerung fur normalwlderstande,

Wiss. Abh. der P.T.R. 24, 51 (1940), 1940, 24, 51.

30. ALFRED SCHULZE, Uber wlderstandswerkstof fe fur normalwldetstande, Wiss. Abh. der

P.T.R. 1940, 24, 91.

31. ALFRED SCHULZE, Uber die verwendung von sllberleglerungen als wlderstandswerkstoffe,

Wiss. Abh. der P.T.R., 1941, 25, 1 1941, 25, 121.

32. New alloy has Improved electrical resistance properties, Materials and Methods,

1948, 28, (No. 2), 62.

33. C. P. MARSDEN, Jr., Three design considerations In selecting resistance alloys,

Elec. Mfg., 1948, 42 (No.2), 116.

(19649)

392
THE ERROR DUE TO THE PELTIER EFFECT IN DIRECT-CURRENT MEASUREMENTS OF

RESISTANCE, C. G. Kirby and M. J. Laubitz (National Research Council of

Canada, Division of Physics, Ottawa, Canada), Metrologia _9, 103-106 (1973).

The recent development of ac bridges for precise resistance

determinations has brought on the not altogether unexpected discovery of

significant differences between dc and ac measurements. In this article, we discuss one such difference, observed in germanium resistance thermometers,

and show that it can be unequivocally ascribed to the Peltier effect, the dc

measurement in this case being in error. This conclusion is based on a thermal model of the resistor which predicts accurately the magnitude and

frequency variation of the observed effect, both for 2- and M-terminal

measurements. The applicability of this model to other devices is briefly

discussed.

RESISTANCE COMPARISONS AT NANOVOLT LEVELS USING AN ISOLATING CURRENT RATIO

GENERATOR, L. Crovini and C. G. M. Kirby (National Research Council of

Canada, Ottawa, Canada), Rev. Sci. Instrum. 4J_, 493-^97, (1970).

A circuit has been developed for the generation of isolated current

having precisely defined ratios. It can be used to compare resistances with

accuracies up to 0.03 ppm and with sensitivities as low as 1 nV. When

comparisons are made using low frequency currents, thermal emf's are eliminated. Several variations of the circuit which can be made easily in

the laboratory are described.

393
I

I
CAPACITORS AND INDUCTORS

Transportable 1000 pF Standard for the NBS

Capacitance Measurement Assurance Program

George Free and Jerome Morrow (1982) 397

Testing to Quantify the Effects of Handling of

Gas Dielectric Standard Capacitors

Charles R. Levy (1 982) 41 2

Losses in Electrode Surface Films in Gas Dielectric

Capacitors

Eddy So and John Q. Shields (1 979) 455

A Programmable Phase-Sensitive Detector for Automatic

Bridge Applications

Robert D. Cutkosky (1978) 461

Measurement of Four-Pair Admittances with Two-Pair

Bridges

John Q. Shields (1 974) 463

Phase Angle Characteristics of Cross Capacitors

John Q. Shields (1 972) 471

Techniques for Comparing Four-Terminal-Pair

Admittance Standards

R. D. Cutkosky (1 970) 475

A Varactor Null Detector for Audio Frequency

Capacitance Bridges

R. D. Cutkosky (1968) 491

Comparison Calibration of Inductive Voltage Dividers

Raymond V. Lisle and Thomas L. Zapf ( 1964) 499

Designs for Temperature and Temperature Gradient

Compensated Capacitors Smaller Than Ten Picofarads

Robert D. Cutkosky (1964) 504

A Technique for Extrapolating the 1 KC Values of

Secondary Capacitance Standards to Higher Frequencies

R. N. Jones (1963) 507

Four-Terminal Pair Networks as Precision Admittance

and Impedance Standards

R. D. Cutkosky (1 963) 521

Some Results on the Cross-Capacitances Per Unit Length

of Cylindrical Three-Terminal Capacitors with Thin

Dielectric Films on Their Electrodes

D. G. Lampard and R. D. Cutkosky (1960) 525

395
A Combined Transformer Bridge for Precise Comparison

of Inductance with Capacitance

Andrzej Muciek (Abstract, 1983)

396
Transportable 1000 pF Standard for the NBS

Capacitance Measurement Assurance Program

George Free and Jerome Morrow

Electrical Measurements and Standards Division

National Bureau of Standards

Washington, DC 20234

/ w \

U.S. DEPARTMENT OF COMMERCE, Malcolm Baldrige, Secretary

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director

Issued October 1982

397
National Bureau of Standards Technical Note 1162

Natl. Bur. Stand. (U.S.), Tech. Note 1162, 15 pages (Oct. 1982)

CODEN: NBTNAE

U.S. GOVERNMENT PRINTING OFFICE

WASHINGTON: 1982

For Sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, D.C. 20402.

Price $2.75

(Add 25 percent for other than U.S. mailing.)

398
TRANSPORTABLE 1000 pF STANDARD FOR THE NBS

CAPACITANCE MEASUREMENT ASSURANCE PROGRAM

George Free and Jerome Morrow

Electrical Measurements and Standards Division

National Bureau of Standards

Washington, DC 20234

Abst ract

A capacitance transport standard has been constructed for

use in the National Bureau of Standards Measurement Assurance

Program. The transport standard was designed so that variations

in ambient temperature and mechanical shock would have a minimal

effect on the value of the internal reference capacitors. A significant improvement in stability of 1000 pF capacitors during

shipment and in the laboratory has been achieved through this

d e s i g n . Key words: calibration; measurement assurance; measurement assurance programs;

reference standards; standard capacitors; standard qualification; transfer

standards

.

Int roduct i on The National Bureau of Standards Measurement Assurance

Program (MAP) in electrical units has been in existence for

several years. In this program, NBS transport standards are

shipped to customer laboratories and calibration of the

customer's reference standards is done in place. The accuracy of

the calibration procedure depends on the degree of knowledge of

the value of the NBS transport standard while it is at the

customer's laboratory; this is optimized by analysis of measurements made at NBS before and after the period of time the

standard is at that laboratory. If the value of the transfer

standard is known accurately, the advantages to the laboratory

are threefold: reduction of down-time due to the absence of

reference standards, greater confidence in the values of

laboratory standards, and documentable verification of the

customer's measurement process through the test procedure.

For the capacitance MAP, it was necessary to develop a transport standard which would behave predictably over an extended period during which the internal capacitors would be

subjected to both mechanical and thermal shock. Due to customer

needs, the transport standard would have to be at the 1000 pF

level (the most common reference standard in most laboratories).

At the 1000 pF level, the most stable, commerciallyavailable standards are nitrogen dielectric capacitors. The

values of these capacitors vary due to temperature (with

coefficients of about 2 ppm/°C), temperature hysteresis, and

mechanical stress. The reduction of instability from these

sources were planned through design of the transport box and

minor modifications to commercially available capacitors.

The goals for the construction of the transport package were

as follows:

399
o The package should provide a transportable temperature

controlled environment with a maximum departure from

nominal temperature of 0.01 °C under normal laboratory

conditions.

o The package should provide for a minimum amount of

change due to handling.

o The package should be of simple design and construction

so that it may be duplicated by laboratories interested

in increasing the stability of their standards.

Construction

The most important limitation in the performance of the

transport standard is the quality of the capacitor used. Tests

were devised by which the' mechanical stability of capacitors

could be systematically evaluated (1) and the results were used

to select the four "best" capacitors available for inclusion in

the standard. The method of testing and the results are

discussed in the Appendix. These tests were also used to

evaluate 10 pF and 100 pF standards.

Before placing the capacitors in the transport box, their

outer enclosures were removed. Since the stability of the

trimmer in these capacitors is unknown and difficult to analyze,

it was also removed and electrical connection to the inner

enclosure was made through a special head. The head was designed

to eliminate stray capacitance between the leads coming out of

the capacitor and to guarantee good electrical contact between

the cable shield and the enclosure, thus making the capacitor more

closely meet the requirements for three-terminal devices with a negligible amount of stray capacitance.

The capacitors were placed in an aluminum box (see Figure 1)

25.4 cm (10 inches) on a side, with sufficient soft-sponge

packing material surrounding them so that they were effectively

fixed in position with no rigid connections. The capacitors were

insulated from each other by plastic sheets, thereby reducing the

possibility of electrical ground loops during measurements.

Heaters, consisting of many turns of magnet wire, were positioned

to cover the total surface of each side of the box and were

attached to the box with adhesive tape (adhesive agent on both

sides). Additional tape (of the type used to seal heating air

ducts) was tightly wrapped on top of the heater wires to assure

good thermal contact with the box and prevent motion of the wire.

An outer box was made out of plastic and wood, its dimensions

allowing for at least 7.6 cm (3 inches) of insulation between it

and the inner box. The aluminum box was then foamed in place

with rigid polyurethane. Coaxial leads from the inner box to the

outside were placed so as to assure good thermal contact with the

inside aluminum box, reducing temperature variations transmitted

through the leads. The coaxial leads are RG316/U 50 ohm cable.

The small cross-section of metal in the inner conductor and

shielding reduces the heat path to the outside.

400
The temperature control circuit for the standard (see Figure

2) consists of a thermistor bridge and a very simple operational

amplifier circuit. The temperature-sensitive components of the

control circuit (resistors and amplifier) are placed inside the

aluminum box in a section separated from the capacitors by an

aluminum plate. The less sensitive components, such as the power

supply and the power transistor for the heating coils, are in a small control box separate from the transport standard. The

control thermistor is wrapped in aluminum foil and placed under

the adhesive tape between the heater wires and the aluminum

enclosure.

A temperature measuring device consisting of a thermistor

bridge with variable decade resistors in one arm was constructed.

The thermistor was placed on top of and taped to one of the

capacitors.

The control circuit was designed to operate from 110 VAC or

a 12 VDC primary power source. When operated from a DC power

source, the circuit needs an additional _+ 1 5 VDC battery to

provide current for the operational amplifier.

Performance

As the standard was too large to fit in available

environmental chambers, the temperature-variation performance

could be studied only with respect to changes in the laboratory

environment. The temperature variation tests were carried out

over a relatively narrow range of temperatures.

For a temperature range of 3.9 °C in the laboratory, i.e.,

from 19.9 °C to 23.8 °C, the measured variation in box

temperature was 0 . 0 1

1

0

C while, for a temperature variation of

1.3 °C from 19.9 °C to 21.2 °C, the measured variation was

0. 005°C. These two tests were carried out in different

laboratory environments.

Tests correlating change in capacitance with temperature

variations were first carried out using the measurement system

described in (1). The standards were intercompared with 100 pF

fused silica dielectric capacitors which were in a temperature

controlled air bath. The small apparent changes in value of the

transport standard could not definitely be attributed to the

transport standard alone due to possibilities of error in the

rest of the system.

The reported measurements of temperature dependence were

obtained using the measurement system described in (2) and 100 pF

quartz dielectric capacitors in an oil bath as the reference

standard. The results of the stability test are shown in Table

1. The period during which measurements were made was one month

(31 days). There were 17 i ntercompari sons and the time-of-day of

measurement was varied in order that the full range of laboratory

temperatures, i.e., 19.9 to 21.1 °C, was included. The

401
COAXIAL LEADS

FIG.1 Schematic diagram of the aluminum box for housing the 1000 pF

transportable standard.

FIG. 2 Schematic diagram of heater control circuit.

402
resolution of the measuring system is +_0 . 0 01 ppm. The range of

capacitance change, C(MAX)-C(MIN), for each standard is given in

the first row of the table.

Table 2 is based upon the same data as Table 1 but the

measured values are represented by the equation y = a + 6t +yAT

using a least squares fit. In this equation y is the departure

from the nominal capacitance value (in ppm) after a time t (in

days), and for the temperature increment AT (degrees celsius)

which is the departure of the box temperature from the nominal

control temperature, or AT = T Rox -30.00 °C. The paramters a, e , and y are, respectively, the residual capacitance offset (ppm),

the daily drift rate (ppm/day), and the temperature coefficient

(ppm/°C).

Comparisons of the residual standard deviation a (RES) with

the standard deviation of an individual measurement a ( I n d . Meas.)

of Table 1 indicates that this equation fits the performance of

the capacitors quite well. The range of temperature variations

of the standards during this test was 0.005 ° C . The similarity

of q (MEAN) in Table 1 and a(RES) in Table 2 indicates that over

a short period of time (one month) the values of the standards

could be calculated either from the mean of a set of measurements

or by using a linear fit.

The methods used to test mechanical stability of individual

capacitors could not be applid to the entire transport standard

because of its size and weight. Mechanical stability was

determined through actual shipment of the box to laboratories.

In this evaluation the capacitors were shipped five times to

various parts of the United States. All snipping was by air

freight with hand carried delivery service to and from the airports. The transport standard was shipped under battery power

for temperature control. No care was taken to ensure that the

shipping container received special handling by the airlines.

With data from shock recorders on similar boxes it was surmised

that the container might well experience forces exceeding 45g

during transport. Table 3 indicates the effects of travel on the

standards

.

The numbers given are absolute values, the sign of the

change being disregarded. Actual changes were both positive and

negative, thus the change in the box mean is generally less than

the changes in individual capacitors. The columns indicate

changes due to one complete round trip. Each change was calculated as Cp-C^ = change, where C

R is the value before shipment

and is the value after shipment. The last row of the table

indicates the changes Cr-C/\» where Cp is the value of each

standard at the start of testing and C A is the value of each

capacitor nine months later. As a general rule, the larger

changes occurred in all capacitors on the same transfer.

403
TABLE 1 Parameter Capacitors

Al A2 A3 A4 Mean

(ppm) (ppm) (ppm) (PPm) (ppm)

Range 0.022 0.080 0.064 0.036 0.046

a ( I n d i vidua! Measurement) 0.009 0.020 0.020 0.01 2 0.01 5

a ( Me a n ) 0.003 0.007 0.007 0.004 0.005

TABLE 2 DAILY VALUES FITTED TO a + st + YA T

Al A2 A3 A4

a(ppm) -1 47.290 -270.1 1 5 -325.593 -1 72.455

6 (ppm/day) -0.0012 -0.0030 -0.0027 -0.0021

Y (ppm/ u C) 3.2667 2.8826 2. 6822 3.8676

a(RES)(ppm) 0.006 0.004 0. 006 0.004

t = time in days AT = T box " 30 ' 00 °

TABLE 3

CHANGES IN CAPACITANCE DUE TO SHIPMENT

Al A2 A3 A4 Mean

(PPm) (ppm) (ppm) (ppm) (PPm)

Min. Change 0.071 0.047 0.030 0.020 0.009

Max. Change 0.388 0. 162 0.209 0.111 0.180

Ave. Change 0.194 0.104 0.083 0.061 0.069

Change for 9 Months 0.123 0.426 0.193 0.081

404
In one shipment, capacitor A3 changed 1.24 ppm. This value

was not included in Table 3, since in a normal MAP procedure,

this change would be sufficient to cause this capacitor to be

excluded from all calculations.

Conclusion

Calibrations using the transport standard appear to offer

significant improvement over the normal method of shipping

capacitors to NBS. At present, a 5-ppm uncertainty in value must

be assigned to capacitors shipped to NBS for calibration; a major

portion of this uncertainty is an allowance for changes due to

shipping. Using the transport standard, this uncertainty can be

reduced by approximately one order of magnitude.

There are several improvements that will be made on future

transport standards. For example, the enclosure is too heavy and

bulky for easy handling. Thus a reduction in both weight and

size will be attempted.

The capacitors used were not initially tested for frequency

response. Since they have varying degrees of frequency

dependence, their range of applicability is limited. Th

conductances of these capacitors are higher than normal

indicating that greater care must be taken in the assembly of the

package on future versions.

The temperature coefficients and drift rates of the

capacitors were calculated for a period of time between

shipments. A further study will be carried out to see if these

coefficients remain unchanged over the time interval of two

measurement periods with shipment of the standards in between.

If the coefficients do remain constant, a further reduction in

the uncertainty of predicted values for the standards will be

justified.

Acknowl edgements

The authors would like to thank Mr. Robert E. Long and Mr.

Charles Levy for their help in the construction and testing of

the transport standard.

References

(1) Cutkosky, Robert D. Capacitance bridge-NBS type 2 . Nat.

Bur. Stand. Report No. 7 1 03 , 1 96 1.

(2) Cutkosky, Robert D. Actual and passive direct reading ratio

sets for comparison of audio-frequency admittances. J. Res.

Nat. Bur. Stand., 6 8 ( 4 ) : 2 2 7 - 2 3 6 ; 1 964 October. Vol. 68C, No.

4, October 1964.

405
Appendi x Capacitors are exposed to a variety of mechanical shocks

during use in the laboratory or during shipment. Great care in

packing standard capacitors will, at most, reduce the intensity

of the shock incurred in shipment; in the laboratory, a capacitor

receives some degree of shock every time it is moved. Capacitors

of the type used in the transport box have parallel plates fixed

in position by support rods. Mechanical shock tends to change

the geometry of the plates or the position of the plates relative

to the encl osu re. The mechanical stability of the standards was of utmost

importance since it was intended that they be shipped from six to

ten times annually. Therefore, six tests of stability were

devised to select those standards most suitable for shipment.

Since only a limited number of capacitors were available as

candidates for testing the transport box, absolute limits of

performance could not be established. Instead, those capacitors

of the available groups which appeared to be "best" were chosen.

The six tests used were designated as ORIENTATION, SMALL

ANGLE, TILT, SOFT KNOCK, HARD KNOCK and DROP TESTS. The test

protocols and the changes of capacitance measured are outlined

below.

ORIENTATION: The capacitor is measured while in an upright

(vertical) position and while in the four posible horizontal

positions. The maximum difference with all measurements

considered is determined. Any change greater than 10 ppm is

usually evidence of a faulty capacitor. This test was used to

determine gross instability.

SMALL ANGLE: The capacitor is measured in the upright

position and on a 3° inclined plane. Four measurements are made

on the inclined plane with the capacitor being rotated 90°

(around the vertical axis) after each measurement. The maximum

difference with all measurements considered is determined. Since

most laboratory tables are not exactly horizontal, the offset

from horizontal would cause error if the capacitor were

excessively sensitive. Differences of several tenths of a ppm

were evident in some capacitors. Only those with a negligible

change were used in the box.

TILT: After measurement of capacitance in the upright

position the capacitor is tilted 90° and returned to the upright

position for the second measurement. The test is repeated for

each side and the bottom of the capacitor, i.e., five tests in

all. The net difference C(UPRIGHT) 1 - C(UPRIGHT) 2 is determined.

Since changes with position should be completely reversible for a good capacitor, any measurable net change was cause for rejection.

SOFT KNOCK: The capacitor is measured before and after

being struck by the head of a pendulum. The head is covered with

406
soft rubber. The force generated by the pendulum is roughly

equivalent to that generated by a mild wrist action motion of the

human hand. The force is in a horizontal plane and it is applied

to the four sides and corners of the capacitor. The differences

C(BEFORE) - C(AFTER) are determined. A minimum change was the

criterion for selection of capacitors from those available as

candi dates

.

HARD KNOCK: The test procedure is the same as that for SOFT

KNOCK with the exception that the soft rubber head is replaced by

a hard plastic head. In the SOFT KNOCK test the force has a relatively long duration with a gauss ian-like shape; in the HARD

KNOCK the force tends to be impulsive with a more abrupt rise and

fal 1 . DROP TEST: The capacitor is measured in the upright position. A

2" plate is inserted under one side of the capacitor raising that

side off the table. The plate is quickly removed allowing the

capacitor to drop back in its initial upright position and the

capacitance is measured again. Differences C ( UPR I GHT)^

C(UPRIGHT)2 are determined for tests repeated for the four sides

of the capacitor. The applied impulse is in the vertical direction. Capacitors were chosen on the basis of minmum change in

this test.

The results of these six tests for a large number of 1 000 pF

capacitors are shown in Figures 3-8. Many capacitors not

specifically considered for the transport standard are included

and, therefore, these graphs indicate the general performance of

the 1000 pF capacitors.

The graphs are histograms for the various tests. The Y axis

gives the percentage of capacitors which fall into specific

ranges of maximum change given by the X axis. The X axis has a limit of either 20 ppm (Figure 3) or 2 ppm (Figures 4-8). The

tests revealed a few capacitors with changes far beyond these

limits. In these cases the position of the mark on the graphs

relative to the limiting value indicate only that they are beyond

the limit and are not intended to show the actual change

measured

.

407
ORIENTATION

PERCENT OF CAPACITORS

VS

MAXIMUM CAPACITANCE CHANGE

T T

^—

V

J=L

2 4 6 8 10 12 14 16 18

CAPACITANCE CHANGE (PPM)

FIG.3 Histogram of the fractional numbers of capacitors for which different values

of maximum capacity change were caused by the Orientation Test.

20 -

LU 15 - o

<

< 10 - CL<

o

5

-

SMALL ANGLE

PERCENT OF CAPACITORS

VS

MAXIMUM CHANGE IN CAPACITANCE

ULn

—i 4=l r 0.2 0.4 0.6 0.8 1.0 1.2 1.4

CAPACITANCE CHANGE (PPM)

FIG. 4 Histogram of the fractional numbers of capacitors for which different values

of maximum capacity change were caused by the Small Angle Test.

408
90 -

80 -

70 - COg 60

CJ 50 - <

Q_

<

CJ 40

O 30

20 -

10

TILT

PERCENT OF CAPACITORS

VS

MAXIMUM CAPACITANCE CHANGE

0.2 0.4 0.6 0.8 1.0 2.0

CAPACITANCE CHANGE (PPM)

FIG. 5 Histogram of the fractional numbers of capacitors for which different values

of capacity change were caused by the Tilt Test

50 -

co 4 o O

CJ<

Q_

<

CJ

U_O

30 -

20

10 -

SOFT KNOCK

PERCENT OF CAPACITORS

VS

MAXIMUM CAPACITANCE CHANGE

I

1 1 1 1—V—i—

0.2 0.4 0.6 0.8 1.0 1.2 2.0

CAPACITANCE CHANGE (PPM)

r~i

FIG. 6 Histogram of the fractional numbers of capacitors for which different values

of maximum capacity change were caused by the Soft Knock Test.

409
50 -

40 H

COO

CJS 30

<

o

20

10 -

HARD KNOCK

PERCENT OF CAPACITORS

VS

MAXIMUM CHANGE IN CAPACITANCE

0.2 0.4 1.2 1.4 1.6 2.0

CAPACITANCE CHANGE (PPM)

FIG. 7 Histogram of the fractional numbers of capacitors for which different values

of maximum capacity change were caused by the Hard Knock Test.

50 -4

co 40

u

<CL 30 <

(J

o^ 20

10 -

DROP

PERCENT OF CAPACITORS

VS

MAXIMUM CHANGE IN CAPACITANCE

T

0.2 0.4 0.6 0.8 1.0 1.2 1.4 CAPACITANCE CHANGE (PPM)

I 1.8 2.0

FIG. 8 Histogram of the fractional numbers of capacitors for which different values

of maximum capacity were caused by the Drop Test.

410
IBS-114A IREV. 2-aci U.S. DEPT. OF COMM. nmi i ap n a nu i o rata BIBLIOGRAPHIC DATA

SHEET (See instructions)

1. PUBLICATION OR

REPORT NO. NBS TN 1162

2. Performing Organ. Report No . 3. Publication Date October 1982

A TlTI C Akin CI IDTITI C 4. 1 1 1 Lt AINU iUD 1 1 1 Lt

Transportable 1000 p

F Standard for the NBS Capacitance Measurement

Assurance Program

5. AUTHOR(S)

George Free and Jerome Morrow

5. PERFORMING ORGANIZATION (If joint or other than NBS, see instructions;

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

WASHINGTON, D.C. 20234

7. Contract/Grant No.

8. Type of Report & Period Covered

Final

3. SPONSORING ORGANIZATION NAME AND COMPLETE ADDRESS (Street. City. State, ZIP) same as #6

.0. SUPPLEMENTARY NOTES

H Document describes a computer program; SF-185, FlPS Software Summary, is attached.

.1. ABSTRACT (A 200-word or less (actual summary of most significant information. If document includes a significant

bibliography or literature survey, mention it here)

A capacitance transport standard has been constructed for use in the National

Bureau of Standards Measurement Assurance Program. The transport standard was designee

so that variations in ambient temperature and mechanical shock would have a minimal

effect on the value of the internal reference capacitors. A significant improvement

in stability of 1000 pF capacitors during shipment and in the laboratory has been

achieved through this design.

L2. KEY WORDS (Six to twelve entries; alphabetical order; capitalize only proper names; and separate key words by semicolons)

calibration; measurement assurance; measurement assurance programs; reference

standards; standard capacitors; standard qualification; transfer standards.

L3. AVAILABILITY

[j^J Unlimited For Official Distribution. Do Not Release to NTIS

fjj] Order From Superintendent of Documents, U.S. Government Printing Office, Washington, D.C.

20402.

3] Order From National Technical Information Service (NTIS), Springfield, VA. 22161

14. NO. OF

PRINTED PAGES

15

15. Price

$2.75

411

USCOMM-DC 6043-P80
Testing to Quantify the Effects of Handling

of Gas Dielectric Standard Capacitors

Electrical Measurement and Standards Division

Center for Absolute Physical Quantities

National Measurement Laboratory

National Bureau of Standards

Washington, DC 20234

U.S. DEPARTMENT OF COMMERCE, Malcolm Baldrige, Secretary

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director

Charles R. Levy

OF

Issued October 1982

412
CONTENTS

1. Abstract

2. Introduction

3. Met hodol ogy ,

4. Equipment Fabricated for Tests

5. Testing Procedure and Results.

6. Connector Test

7. Orientation Test

8. Small Angle Test

9. Tilt Test

10. Knock Soft Test

11. Knock Hard Test

12. Drop Test

13. General Considerations

14. Data Reduction

15. Unusual Problems

16. Conclusion

17. Additional Observations

18. References

19. Appendix 1

20. Appendix 2

413
LIST OF FIGURES

Figure 1. Tilting table for Angle Test

Figure 2, Special fixture for Knock Test

Figure 3. Reference points - all tests

Figure 4.

Fi gu re 5.

Figure 6Angle Test - special block for 3 degrees in place .

Figure 7. Tilt Test in progress - side A

Figure 8,

Figure 9. Set-up for simulated Drop Test

ACKNOWLEDGMENT

I acknowledge with appreciation Dr. Arthur 0. McCoubrey

for his expertise, time, and effort in the editing of this

paper.

414
TESTING TO QUANTIFY THE EFFECTS OF HANDLING

OF

GAS DIELECTRIC STANDARD CAPACITORS

by Charles R. Levy

1. Abstract

A test method known as the "Handling and Stability Test" is

currently being used at NBS as part of the requirement for the

five-part-per-mi 1 1 i on calibration of gas dielectric standard

capacitors. This test is used to achieve qualitative information

on the effects of mechanical shock from shipping and handling on standard capacitors and to rank them quantitatively with respect

to these effects.

2. Introduct i on

Nitrogen dielectric capacitors are used as the reference

standards for measurements of capacitance in many metrology

laboratories in the United States, as well as in many other parts

of the world. Since the most common standard used has a capacity

of 1000 picofarads, this report is based on tests of capacitors

of this value, but an equivalent test is performed on 10 and 100

pF capacitors as well.

Recognizing the range of mechanical shocks which these

capacitors may receive in service or in transport, NBS devised

tests to confirm the mechanical integrity of the units before

calibration and to determine the magnitude of any related effects

[1]. Such effects may then be taken into account quantitatively

in the systematic error portion of the uncertainty statement.

The results of the handling test were also expected to be useful

to guide the user concerning changes in capacitance to be

expected as a result of transport and normal handling. The use

of standard gas filled capacitors in transport service is rapidly

becoming a more important basis for the i ntercompari son of

reference standards among calibration laboratories. A similar

approach, based upon the use of a compressed gas capacitor

(specifically designed for high voltage) in transport service,

has also been investigated in connection with the i ntercompa r i son

of high voltage capacitance measurements [2].

The test procedure has also

a very stable, transportabl

particular, the results of these

of capacitors and in the design

shipping the capacitors.

been used for the development of

e capacitance standard. In

tests were used in the selection

of a suitable transport box for

If the procedures described within this paper are used, gas

dielectric capacitors not normally shipped to a primary reference

facility may be tested to identify possible handling problems

that may not otherwise be detected. In some metrology

laboratories, step-up or step-down methods are used (3, 4, 6, 7,

8) to compare capacitors having a range of value with a high

415
quality (e.g., fused silica) reference capacitor. In such cases,

the uncertainties assigned to the measurements should include

appropriate allowances for capacitance changes associated with

normal laboratory handling. The handling test procedures are an effective tool for this purpose and, in those laboratories having

the necessary facilities and data analysis skills, there is a significant possibility for reducing the cost of traceability to

primary standards. Such cost savings may result from a reduction

in the number of reference standards in inventory to replace

those taken out of service for calibration at a remote facility,

a reduction of the cost of a larger number of such calibrations,

and a reduction in the risks of shipping fragile devices.

3. Met hodol ogy

Tests simulating the acceleration forces during various

types of handling had to be defined. Some guidance in this

regard was obtained from the manufacturers' specifications. On

the basis of these tests, specialized equipment, relatively simple in design, was developed by Jerome Morrow and George Free of

NBS [1]. This equipment and the procedures used to affect these

tests are described in later portions of this document. In

addition, dimensions of the equipment are given in Appendix 2 as

an aid to those who wish to duplicate it.

The tests which have been developed address many of the

common handling problems users may encounter. The test names and

their purposes are:

o CONNECTOR TEST - Determines looseness of either or both

connectors and associated parts. Also used to determine

conductance problems that may be due to poor bonding of the

connector to the case of the capacitor standard, poor solder

joints, or loose inner pins of the connectors.

o ORIENTATION TEST - Used to determine the change in

capacitance due to a ninety-degree rotation of the capacitor

from its upright position; i.e., to a position of lying on

its side. The capacitor is tested six times, once to each

side. Excessive changes may indicate looseness of fasteners

within the capacitor housing.

o ANGLE 3 DEGREES - Most lab table surfaces vary from the

horizontal and this test was planned [1] on the basis of the

estimated worst case in normal calibration laboratory

operations considering a survey within NBS. The value of

three degrees does not reflect a sampling of other

laboratories. The change in measured capacitance allows the

user to relate errors to surface differences of lab tables.

o TILT TEST - This test was devised to check for movement of

component parts within the capacitor housing that may not be

detected by measurement of the standard capacitor in its

normal resting position.

416
417
418
o KNOCK SOFT OR HARD TEST - To simulate impacts the standard

capacitor receives while in transit or through rough

handling in the lab.

0 DROP TEST - Designed to simulate the capacitor standard

being dropped during transport or excessively rough handling

within the users' lab or shipping facility.

4. Eg u i pmen t Fabri cated for Tests

Special equipment was constructed at NBS to accomplish the

tests described in this paper. For the convenience of those

interested in reproducing this equipment, drawings are included

f ol 1 owi ng the text

.

For the angle test, a tilting table was built consisting of

a platform supporting a hinged plate that can be tilted in half

degree increments. Three adjustable feet are used to level the

platform. To adjust the angle of the tilting plate, an

adjustable screw rod with a large knob is provided as shown in

figure 1. This knob can vary the angle of the adjustable plate

from zero to three degrees or more as required. However, as an

angle of three degrees was selected as the most useful test angle

and to save time in adjustment of the elevated table, a removable

block with an adjustable screw was fashioned and fixed in the

three degree position.

For the knock tests, a stand was made to support a hammer

with interchangeable heads. Referring to figure 2 , an angle

plate on the side of the stand was marked off in increments of

1 0, 1 5 , 20, 2 5 , and 45 degrees. A stop is used to prevent the

hammer from going back further than the desired angle. The plate

is movable in the vertical plane to accommodate taller capacitors

without affecting the desired striking angle.

The hammer selected was manufactured by the Nicholson File

Co., Providence, R. I., and is designated as No. 155. The two

detachable heads used are manufactured by J. H. Williams and

Company, United Greenfield Division of TRW, Inc., Buffalo, N. Y.;

they are designated as HSF-15-SS (soft head) and HSF-15-N (hard

plastic head) 1

.

In the case of the drop test, a block of wood having dimensions 3.8 cm (1.5 inches) by 8.9 cm (3.5 inches) was used. By

inserting this block with the smaller dimension under one of the

capacitor sides and then quickly removing it, one drops the

capacitor to that side. This test simulates dropping a capacitor

contained within an appropriate shipping container on one of its

sides or corners.

Certain commercial equipment, instruments, or materials are identified in this

paper in order to adequately specify the experimental procedure. Such identification does not imply recommendation or endorsement by the National Bureau of

Standards, nor does it imply that the materials or equipment identified are necessarily the best available for the purpose.

419
5. Test i ng Procedure and Results

To produce uniform results, a systematic method of testing

was developed. The results collected over the years have, therefore, consistent significance and may be examined for indications

of impending problems, or to predict the direction in which the

value of the capacitor may change. They also provide a history

of the handling and stability of the standard. Furthermore, the

test results can be used to screen capacitors not suitable for

precision measurements and to select capacitors for a bank.

A long set of coaxial leads was connected to the test

capacitor and to the appropriate transformer tap on the NBS-TYPE

2 Bridge (8, 9). A known capacitance standard was connected to

the appropriate tap on the standard side of the bridge as outlined in Table 1.

Table 1. Connection of test unit (unknown) and standard

reference capacitor to NBS-type 2 bridge

Test Capacitor Standa rd Capaci tor

Nominal Bridge Transformer Nominal Bridge Transformer

Val ue Tap Val ue Tap

1000 pF + 1 Fx 100 pF - 1 Fs

100 pF + 1 Fx 100 pF - 1 Fs

10 pF +1 Fx 10 pF - 1 Fs

The procedure adopted at NBS provided for testing the

capacitors in a specific order: connector, orientation, angle

three degrees, tilt, knock soft, knock hard, and finally the drop

test (1). The detailed description of each is given later in

this paper.

A uniform method of identifying the sides of the capacitor

was established as specified in figure 3. Throughout this

report, mention will be made to the reference position.

Referring to figure 3, the reference position in the case of all

tests except the knock test is that position of the capacitance

standard resting upright on the lab table with Side A facing the

operator. In the case of the knock tests, Side A faces the

striking force (refer also to figure 2).

The long coaxial leads to the capacitor are used to allow

for complete movement of the capacitor within the testing area as needed. The capacitor in all tests is connected to the bridge

with the capacitance standard selected using Table 1 and in

accordance with NBS procedures outlined in references (3), (4),

and (5). All measurements start with the capacitor in the

reference position and a return to the reference position after

all movements associated with a particular test are accomplished.

420
421
The long leads to the unit under test do not affect the accuracy

of these measurements because only changes from the initial

values are significant.

6. Connector Test

This test was devised to detect several common problems

experienced with capacitors received for calibration at the NBS.

Such problems are not necessarily related to transportation

effects. Many standards arrive with loose connectors, loose

connector inner conductors, or broken or damaged connector parts;

for example, a leaf may be missing from the inner conductor of

the GenRad 874 connectors, those most commonly used.

Repairs made at the user's lab have been found to create

significant problems due, particularly, to improper soldering

techniques. Unacceptably high or unstable conductances are often

the result of cracked or cold solder joints at the inner pins of

the connectors or at points of connection to the capacitor;

likewise such faults may be associated with a loose inner conductor in either or both connectors. Modifications of the unit

often lead to similar problems (6).

The procedure to determine if a connector defect exists is

to apply a gentle force in a rotary fashion to each lead connector. The bridge must be balanced prior to this test. Any

changes in capacitance or conductance are noted. Then, with the

source oscillator voltage to the bridge off (to avoid transients

in the bridge circuits), each connector is disconnected and then

reconnected. Repeat, recording any difference observed. Changes

greater than 0.05 ppm capacitance or 10 micromhos conductance

are cause for concern. If a problem is apparent, check the

points mentioned above to see if improvement can be achieved. In

some cases, the rotation procedure will improve the readings

without doing anything to the connector; this suggests dirt or

oxidation on the connector surfaces or one of the other problems

mentioned earlier. In such cases, the test should be repeated in an effort to obtain consistent results.

7. Orientation Test

This test is a good indicator of looseness in the web

assembly (refer to figure 4), the case components, and possible

problems within the sealed capacitor assembly.

The most frequent fault detected is a loose web assembly

(refer to figure 4). Web assembly screws loose by one or more

turns have been found to cause changes of several tenths of a part per million. Defects within the sealed capacitor, such as

strains and mechanical displacement of the capacitor plates, may

also be detected (6).

The procedure for this test is quite simple. The test

capacitor is first placed with Side A facing the operator (refer

to figure 3) and the capacitance bridge is balanced. This is the

422
423
reference position. Then the capacitor is moved from the upright

position to a position of rest on Side A (refer to figure 5), the

bridge rebalanced, and the capacitance value recorded. Without

bringing the capacitor to the upright position, it is turned

successively to positions of rest on Sides B, C, and D, the

capacitance value for each position being recorded. After

capacitance values for positions on all sides have been measured,

the capacitor is returned to the upright position facing Side A

(reference position) and a final measurement is made. The last

reading, identified as ref2, is used to check the firs't (refl)

for drift (ref 1-ref 2=dri ft). Should the difference between the

reference value and the value associated with any position be

greater than ten ppm, the test should be repeated to determine

whether or not the chahge is consistent or erratic. Two or more

tests are desirable to reduce the possibility of a reading or

recording error. Always use the last reference point in a series

of measurements. If the capacitance value changes are not

consistent from test to test, it is advisable to check the

capacitor for defects such as loose screws, nuts, etc.,

including, in particular, those associated with the trimmer

capacitor contained inside the outer enclosure and adjacent to

the sealed capacitor assembly.

As indicated earlier, capacitors supplied from two different

sources were evaluated. In the case of the models involved, the

manufacturer's published specifications for maximum capacity

changes under the conditions of the tests ranged from 1 ppm to 10

ppm. Changes greater than manufacturer's published specifications were considered as evidence of a defective capacitor.

8. Smal 1 Angl e Test

This test was originally done for changes from zero to one,

zero to two, and zero to three degrees. However, this series of

tests was time consuming, and a single test for a change from

zero to three degrees was found to be just as useful. The angle

of three degrees was selected as the worst case by measuring a number of laboratory tables and finding that surfaces are usually

within plus or minus two degrees of the horizontal. The change

in capacitance due to small changes in angle of tilt is generally

found to be a linear function of the angle within the zero to

three degree limits. A worst-case acceptable value of 0.2 ppm

per degree of angle change has been established for capacitors in

satisfactory condition [1].

The procedure for this test is not difficult. However, the

placement of the capacitor, adjustment of the elevated table, and

provisions to avoid movement of the laboratory table or the

elevated table are all important for consistent results (refer to

figure 6). Be sure the adjustable table is horizontal (zero

degrees). Then place the capacitor in the center of the tiltable

plate on the adjustable table upright with Side A facing the

operator. The measured capacitance under these conditions becomes the reference value and should be recorded as such. Use the

adjustable knob to increase the angle to three degrees or, alter424
425
nately, use the special block made for three degrees, located

securely in a suitable slot to avoid error in the elevation of

the table. (Refer to figure 6). Measure the capacitance values

with Sides A, B, C, and D facing the operator. Differences

greater than 0.2 ppm between the reference value and the value

associated with any other position indicates that the test should

be repeated. If the results of a second series of measurements

are similar to those of the first run, the changes probably

reflect a characteristic of the particular capacitor. If the

test is repeated, and the results are significantly different

from those of the first test, subtract the smallest change from

the largest change and record one half the result; this value

should not be greater than 0.2 ppm. If it is larger, check for

looseness in the web assembly.

9. Tilt Test

The tilt test (rotation through angles of 90 degrees, 180

degrees and return) serves several purposes. The geometry of

parallel plate capacitors is such that movement of the plates

will cause changes in capacitance. For example, shifting of the

capacitor plates by only a few thousandths of an inch will cause

changes ranging from a few tenths of a part per million to

several parts per million. If all accessible fasteners holding

the capacitor to the web assembly are tight and a sudden change

in capacitance can be detected upon tilting and it does not

disappear within plus or minus 0.1 ppm when the capacitor is

placed back in the upright position on the table, it may be

assumed that a problem exists within the sealed capacitor

assembly

The testing procedure is as follows. First, place the

capacitor in the reference position on the table with leads

connected as in previous tests. Balance the capacitance bridge

as usual. Then grasp the handle of the capacitor and by

balancing the weight of the capacitor in the other hand (refer to

figure 7), turn (gently) the capacitor to a position with Side A

in the horizontal plane. Note any erratic changes of the

detector reading as you place the capacitor in the upright position on the table. Record any change observed. Repeat the test

for Sides B, C, and D. Finally, repeat the test for the inverted

position (refer to Figure 8). Grasp the capacitor and turn it

upside down again observing the detector for changes as you

return the capacitor to the normal orientation and place it back

on the tabl e. Changes greater than 0.2 ppm between any two measurements of

the capacitance of the upright capacitor indicate the possibility

of a problem and the test should be repeated at least once. If

the change is greater than 0.2 ppm, and consistent with the

previous test, the test may be considered to reflect a characteristic of the capacitor. If changes during a second test

are significant and do not repeat the results of the first test,

check for looseness in the web assembly and capacitor assembly

mounting parts. In the worst case, you might hear a distinct

426
427
Figure 7. Tilt test in progress - side A horizontal.

428
Figure 8. Tilt test in progress - test unit inverted.

429
"clunk" as you invert the capacitor indicating that the capacitor

assembly within the case will probably be found to be loose.

Visual observation of the detector during this test is

essential. Erratic behavior, indicated by a sudden change in the

detector zero point, may be evidence of problems such as movement

within the capacitor. This should be recorded to determine, in

the future, if the quality of the capacitor degrades with time.

It is also suggested that such a capacitor be calibrated more

frequently as behavior of this nature is generally a sign of

impending deterioration. Before reaching a conclusion that the

capacitor is faulty, be sure to check all fasteners previously

mentioned as well as connection points and cables. In many

cases, a correctable fault can be eliminated.

10. Knock Soft Test

The mechanical shock in the knock soft test is approximately

equal to a mild, wrist-action motion of one's hand against the

side of the capacitor. The test is a controlled impact of a soft

ru b be r -t i p ped hammer swung, in the vertical plane, from a position of 45 degrees. The hammer strikes the test unit on a side, just below the horizontal plane of the top surface, and

midway between adjacent sides.

This test may reveal several conditions. For example, defects within the sealed capacitor enclosure, or looseness of

fasteners in the web assembly of the capacitor, may be

responsible for shock -i nduced changes of capacitance.

During transit, the capacitor is generally subjected to

larger mechanical forces than this test will produce. However,

small impulsive forces often create more noticeable changes

within a capacitor than larger, less impulsive forces. This test

is intended to simulate a typical impulsive force that a wellpacked capacitor may receive during handling in transit. There

are no specified manufacturers' limits for such shocks.

For the test, use the stand with the soft tip end of the

hammer head toward the face of Side A of the capacitor. Balance

the bridge and record the reading. This capacitance value is the

reference value. The hammer is then drawn back to the 45 degree

position and released to strike the capacitor (refer to Appendix

2 , Figures 2 and 4). However, it is not allowed to strike a second time following rebound. Again the capacitance value is

recorded. The capacitor is moved to position A Corner (refer to

figure 3) for a hammer strike. The measurement is repeated

through all sides and corners in the order: Side A, Corner A,

Side B, Corner B, Side C, Corner C, Side D, Corner D.

If changes are greater than 0.2 ppm between any two tests in

this series, it is suggested that the series be repeated. If the

relative changes between a first series and a second series is

noted to be approximately the same, subsequent tests are not

neces sa ry

430
11. Knock Hard Test

This test is similar to the knock soft test. The

differences are that the hammer swing is limited to fifteen

degrees and the hard tipped head strikes the capacitor. The

object of this test is to simulate shocks received by the test

capacitor when, in being lifted from a work bench, it accidentally strikes another object. Mechanical shock in transit may,

under some conditions, be similar to this type of shock. This

test is particularly effective for detecting problems associated

with the trimmer capacitor.

Changes larger than 0.2 ppm should be recorded, particularly

when two sets of readings do not repeat. Changes greater than

0.5 ppm are cause for concern and the capacitor should be

carefully checked for correctable defects. If there are no loose

components, the excesssive capacitance change may be a result of

stress changes within the sealed capacitor enclosure.

12. Drop Test

The drop test was devised to simulate a shock which may

occur in service, for example, due to a drop from the hand to a table surface. It may also simulate, to some extent, the dropping of the standard in its shipping container during handling in

transit. Short-duration forces in the vertical direction may

cause various movements of plates or support rods. Intermittent

electrical shorts between plates may sometimes be found using

this method.

Testing the unknown capacitor requires a block of wood

having measurements of 3.8 cm (1.5 inches) x 8.9 cm (3.5 inches)

and from 10 to 15 cm (4-6 inches) long (refer to figure 9).

Balance the capacitance bridge with the capacitor normally

resting on the table, being sure that all four feet of the

capacitor are in contact with table surface. Then place the

block of wood under Side A (smallest dimension vertical) and

carefully, but swiftly, pull out the block. After the block of

wood has been removed and the capacitor has struck the table,

record any changes in capacitance value. The test is repeated

raising each side of the capacitor sequentially (Side A, B, C,

and D). Changes greater than 0.2 ppm between any two capacitance

values, or between the reference value and any other value should

be recorded. Changes greater than 1 ppm indicate the possibility

of a problem capacitor.

431
432
13. General Considerations

All of the tests described in this document should be

performed in a systematic manner. No test should be discontinued

until all specified sides, corners, etc. have been included. The

specified order of tests, connectors, orientation, small angle,

tilt, knock soft, knock hard, and drop test should be followed to

eliminate possible confusion in comparing results with those of

earlier or later tests. This order was determined to give the

most reproducible results. There is one caution that should be

mentioned. If any fasteners are suspected of being loose, either

tighten all fasteners before the handling test is undertaken, or

wait until all tests are completed. Otherwise, it will not be

possible to know if the repairs caused or cured the problem(s).

It is not necessary to complete a series of tests in one

day; it is only important that the tests be done in the proper

sequence. This is the only way to obtain a repeatable characterization of the standard.

Capacitors determined to be faulty should be retested before

any repairs are attempted in order to eliminate the possibility

of other factors such as improper hook-up, faulty electrical

leads, bridge problems, or faulty connections within the test

system being used.

The use of two reference standards, having known characteristics, to which the unit under test is compared, one during the

first half of the test series and the other during the second

half, may help reveal problems external to the test capacitor.

Such problems might include intermittent faults in a reference

capacitor, improper hook-up, bridge malfunction, faulty leads, or

faulty connectors.

14. Data Reduction

The data for any of the test series described in the

previous sections of this document may be represented by a string

as foil ows :

ref, rdg

1 rdg2 rd 9 n

> where r e f = initial reference value which appears only at the

beginning of a string for all positions to be tested, and

rdg.j =readi ng for a particular side or corner.

Successive differences or changes would then be as follows:

433
d i f , == r d n .

' u y \

ppf .

di f

2

== rdg

2

• rdg

x d1f

3

== rdg

3

- rdg

2

dif n = rdg^rdg,.^

where dif = difference, rdg^ = reading at position i, and ref = reference

The range between any two values is used for two reasons.

In the first place, most laboratories only need to know the

maximum change expected. In the second place, if changes were

nearly equal but opposite in sign, the average would

underestimate possible changes. Changes noted for each side or corner in a string usually will not be randomly distributed,

particularly when significant changes are experienced.

For the tests listed, the following data should be obtained:

ORIENTATION: Minimum of six data points required - refl, Side A,

Side B, Side C, Side D, ref2

ANGLE 3 DEGREES: Minimum of six data points required - ref i , Side A, Side B, Side C, Side D, ref

2

TILT: Minimum of six data points required - ref, side A, side B,

Side C, Side D, upside down, rightside up

KNOCK SOFT OR HARD TEST: Minimum of nine data points required - ref, Side A, Corner A, Side B, Corner B, Side C, Corner C, Side

D, Corner D

In the orientation and angle tests, drift can be determined

from the difference between the first reference and last

reference in a given string. If the test is completed with

minimal time between measurements (less than a half hour), drift

in excess of 0.2 ppm should be recorded. Most excessive drift

will be observed during the first few minutes of testing. In

some cases, drift may continue for several minutes or as long as

a half hour. Such drift may be caused by changes in temperature

of the capacitor under test or changes in temperature of the

standard used as the reference. While the latter may be placed

in a lag box to reduce the effects of rapid temperature

fluctuations, it may not be practical to do so for the capacitor

434
being tested. Therefore, these tests should each be carried out

rapidly to minimize temperature-created anomalies in the data

(which one would have a difficult time sorting out from the

effects of defects) and take place in a relatively wellcontrol led room.

Except for the results of the orientation test, the magnitude of changes caused by handling tests should be included in

the uncertainty of calibration results as a systematic error.

This is particularly important if any of the changes are greater

than 0.5 ppm. In the case of capacitance standards which are not

kept continuously within the metrology laboratory or are otherwise exposed to unusual handling conditions, experience at NBS

suggests that the calibration uncertainty should be increased by

the addition of half the largest change induced by the handling

tests described in this document [1].

The stresses described in this document are believed to be

small compared to those that are encountered in normal shipping

and handling; therefore, unusual effects due to shipping should

be determined by using the results of control measurements made

in the user laboratory before and after shipping in addition to

the results of these tests at NBS.

15. Unusual Probl ems

Gas dielectric capacitors have two sources of capacitance

change which are not repairable by the user; in the first place,

all gas dielectric capacitors manufactured in the United States

use a trimmer capacitor to bring the value of the capacitor close

to its nominal value. Many users adjust this capacitor to meet

specific needs. Two types of trimmers are in use, one having an adjustable core and the other having interleaved plates similar

to a radio tuning capacitor. The plate type is rarely found to

be a problem unless it becomes loose. Since this type is

generally exposed within the capacitor housing, a loose part,

such as a nut or screw, falling between the blades of the trimmer

capacitor may cause an electrical short or mechanical damage to

the plates. Such faults may cause capacitance changes of greater

than 100 ppm. Changes in placement of the leads to the trimmer

may also affect the capacitance by several ppm in either

direction without any change in the trimmer adjustment.

The coaxial adjustable-core type of capacitor trimmer has

been known to short, or worse, fall apart due to extreme abuse in

transit. If there is a possibility of an electrical short, the

trimmer must be replaced because excessive current will be drawn

through the measuring system. Malfunction of the trimmer may

cause a capacitance change of several hundred ppm.

Removal of either type of trimmer will result in capacitance

changes of up to 300 ppm (determined for the capacitors selected

for the NBS portable transport box). If such departures from

435
nominal value are acceptable, it is suggested that the trimmer

capacitor be removed to eliminate a possible cause of

i nstabi 1 i ty

A second, less frequent, problem involves hermetic seal

leakage. The hermetic seals of capacitors manufactured in the

United States are generally trouble free unless undue pressure is

applied on the electrical feed-through connection points or they

are overheated. Cracked seals are not common; however, if this

is suspected, temperature cycling will reveal the fault. Cooling

to freezing followed by warming to approximately forty degrees

Celsius and returning to lab temperature should be sufficient to

verify seal leakage. A capacitor that is cycled will not retrace

to the previous value within the tolerance stated by the manufacturer unless the seals are intact. There may be problems within

the sealed capacitor other than those associated with the seals.

However, any capacitor that cannot be cycled without significant

changes should not be considered satisfactory as a working

standard

16. Cone! usi on

A review of the graphs for the various test procedures will

disclose the utility of handling tests described in this

document. The overall results of a number of such tests

conducted at NBS are summarized in table 2, following:

Table 2. Summation of tests

TEST ACCEPTABLE LIMITS* (PPM) % FAILED

Orientation 10 23

Angle 3 Degrees 0.2 48

Tilt 0.2 5

Knock Soft 0.2 14

Knock Hard 0.2 13

Drop 0.2 14

Acceptable limits excluding orientation test for which one half

the largest change in any test should be added to the uncertainty

of calibration.

The tables and graphs (Appendix 1) in this document are

derived from the results of tests carried out on 200 capacitors

during the period 1972-7 8. The method of evaluation involved

collection of data from all capacitors received for testing the

first time in order to establish a basis unbiased by the selection criteria and to reflect a good cross-section of all

capacitors received. Data are selected randomly from all

436
capacitors received for test in order to avoid disclosure of

differences between the products of different manufacturers.

Data from succeeding years were also selected randomly for

capacitors not used in previous data evaluations in order to

obtain an overall balance.

Data from the first year of testing were compared with data

from succeeding years with very convincing results. In more than

80% of the cases compared, the results of the capacitors repeated

to within 0.2 ppm of those from the previous year for each

particular test. Most capacitors with changes greater than 0.2

ppm had been subjected to rough treatment. For fewer than 10%

there was no discernible reason for the changes noted.

Referring to the summary of tests, it is apparent that the

orientation and three degree angle tests were failed more often

than the other tests. If a capacitor shows poor results on the

tilt test, the problem can, in principle, be overcome by using a level table. In the case of the orientation problem, there is

not much that can be done other than use the capacitor in the

upright position. It should be recognized that excessive changes

during these tests suggest a lack of mechanical integrity within

the standard capacitor and reliabilty during service is

open to question.

It will be noted that there is no graph in Appendix 1 for

the connector test. Generally, the changes during this test are

small and when a problem with a connector is corrected, there is

generally no remaining concern.

17. Additional Observations

The care with which a gas dielectric capacitor is packed for

shipping can, to a large extent, determine its useful life.

Manufacturers in the United States ship standard capacitors in

protective reusable containers, which should also be placed in an outer box containing at least 10 cm (4 inches) of shock -absorbi ng

material on all sides for added protection.

Exposure of the capacitor to abnormal temperatures, such as those found in the trunk of a car on a hot day, should be

avoided. The time for the capacitor to stabilize after being

exposed to excessively low or high temperatures depends on the

design of the capacitor. A similar consideration applies

following a long period of storage with the capacitor resting on

a side; a period of recovery in the normal upright position is

necessary before its use or before calibration at NBS.

Checking all screws, nuts, etc., for looseness before the

capacitor is shipped for calibration will reduce the possibility

of damage due to the movement of parts during transit. Loose

connectors should be properly positioned and tightened before

tests are started.

437
Handling the capacitor in the laboratory or field is as

important as care in shipping if long term stability is

essential. Capacitors receiving rough treatment, such as impacts

against other equipment or furniture, may be damaged in terms of

loosened parts or more serious mechanical changes.

It is important to document values and abnormal changes

found in the handling tests. Both can guide the user in avoiding

problems and removing from service a faulty capacitor; many hours

of needless measurements can, thereby, be saved.

All maintenance and repairs should be performed after

completing the series of tests described in this document unless

a definite problem is detected. In this way, problems discovered

during tests may be evaluated as correctable or otherwise

identified as an intrinsic characteristic of the unit under test.

The material contained within this paper is intended to

serve as a guide; it is not suggested that the problems discussed

are the only problems the user will experience. However, the

handling tests described have been found to disclose the most

frequent problems encountered at NBS. All known causes of

capacitance change should be documented fully and the capacitor

should be monitored systematically if it is to be used at the

five ppm level of uncertainty.

18. References

[1] Free, G. and Morrow, J., Transportable 1000 pF capacitance

standard. Nat. Bur. Stand. (U.S.) Tech. Note 1162; 1 982.

[2] Anderson, W. E., Davis, R.S., Petersens, 0., and Moore, W.

J. M., An International Comparison of High Voltage Capacitor

Calibrtions, IEEE Trans. Power Appar. Syst. PAS-97(4) , pp.

1217-1223 (1978).

[3] Thompson, A.M., The precise measurement of small

capacitances. IRE Trans. Instrum. 1-7(3-4) : 245- 25 5 ; 1958

Dec.

[4] Cutkosky, R.D., Four-terminal pair network as precision

admittance and impedance standards. IEEE Trns. Commun.

Electron. 70:19-22, 1964 Jan.

[5] Homan, D. N., Application of coaxial chokes to a-c bridge

circuits. J. Res. Nat. Bur. Stand. 7 2C ( 2

) : 1 6 1 - 1 6 5 ; 1968

Apri 1 -June.

[6] A highly stable reference standard capacitor. GR

Experimenter. 37(8); 1963 August.

438
Thoma, P., Absolute ca 1 ori met ri c determination of dielectric

loss factors at = 10 s" 1 and application to measurement of

loss factors of standard capacitors at room temperature.

IEEE Trans. Instrum. Meas. IM-29(4); 1980 Dec.

McGregor, M. C • , Hersh, J. F . , Cutkosky, R.D., Harris, F.

K., and Kotter, F. R., New apparatus at NBS for absolute

capacitance measurements. IRE Trans. Instrum. 1-7:253-261;

1958 Dec.

439
Appendi x 1

Histograms of the Effects of Handling

and Stability Tests on Standard Capacitors

The following histograms are based upon the results of

performing each of the Handling and Stability Tests on each

capacitor of the test group. The maximum capacitance change

value caused by a given test performed a number of times on each

capacitor was used to determine the numbers of capacitors for

which such change values fell within the ranges indicated in the

hi stograms

440
441
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OLl. DZhHW HUJWhUJQ ? • °A 3— a co < >
aoo

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445
en

ui

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Append i

x

2

Drawings of Test Apparatus

The following drawings represent the apparatus used at the

National Bureau of Standards for the tests described in this

publication. Dimensions are given in metric units and customary

units used in the United States.

447
Appendix 2, Figure 1.

KNOCK STAND ANGLE PLATE PATTERN

SCALE I I I I I I I I I I I 0 1 2 3 4 5 cm.

I_J L_J I I I I I 0 1 2 Inches

448
Appendix 2, Figure 2.

Hammer for use with Knock Stand (see following figure)

grams lbs.

HARD RUBBER HEAD 41.03 .09

SOFT RUBBER HEAD 46.70 .10

HAMMER-HEADS 441.14 .97

TOTAL Wt. OF HAMMER 528.87 1.16

(10.63 cm)

449
450
451
452
453
NBS-114A (rev. 2-8C) U.S. DEPT. OF COMM.

BIBLIOGRAPHIC DATA

SHEET (See instructions)

1. PUBLICATION OR

REPORT NO.

NBS TN 1161

2. Performing Organ. Report No.J 3. Publication Date October 1982

4. TITLE AND SUBTITLE

Testing to Quantify the Effects of Handling of Gas Dielectric Standard Capacitors

5. AUTHOR(S)

Charles R. Levy

6. PERFORMING ORGANIZATION (if joint or other than NBS. see instructions;

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

WASHINGTON, D.C. 20234

7. Contract/Grant No.

8. Type of Report & Period Covered

9. SPONSORING ORGANIZATION NAME AND COMPLETE ADDRESS (Street. City. State. ZIP)

same as #6

10. SUPPLEMENTARY NOTES

^} Document describes a computer program; SF-185, FlPS Software Summary, is attached.

11. ABSTRACT (A 200-word or less factual summary of most significant information.

bi bi iography or literature survey, mention it here)

If document includes a significant

A test method known as the "Handling and Stability Test" is currently being used at

NBS as part of the requirement for the five-part-per-million calibration of gas-dielectric standard capacitors. This test is used to achieve qualitative information on the

effects of mechanical shock from shipping and handling on standard capacitors and to

rank them quantitatively with respect to these effects.

12. KEY WORDS (Six to twelve entries; alphabetical order; capitalize only proper names; and separate key words by semicolons)

calibration; measurement assurance; reference standards; standard capacitors;

standard qualification.

13. AVAILABILITY

Unlimited

| | For Official Distribution. Do Not Release to NTIS

£>2 Order From Superintendent of Documents, U.S. Government Printing Office, Washington, D.C.

20402.

3] Order From National Technical Information Service (NTIS), Springfield, VA. 22161

14. NO. OF

PRINTED PAGES

44

15. Price

$4.75

454
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-28, NO. 4, DECEMBER 1979 Losses in Electrode Surface Films

in Gas Dielectric Capacitors

EDDY SO, MEMBER, IEEE, AND JOHN Q. SHIELDS

Abstract The surface characteristic of a number of electrode materials including stainless steel, brass, brass with various plated

surfaces, aluminum, and anodized aluminum has been examined over

a wide range of humidity. The measurements of surface characteristic, expressed in microradian millimeters, have an estimated

uncertainty of 0.01 /irad - mm

I. Introduction

ONE STEP in studying the properties of a dielectric material is to use it as the dielectric of a capacitor and

to determine its loss angle relative to a reference capacitor.

Thus the behavior of the reference capacitor provides the bound of the measurement; and it becomes important to

establish the limits within which its loss angle can be

specified. In the absence of solid dielectric from the direct capacitance of a parallel-plate guard-ring capacitor, its loss

is a function of its electrode-surface films and of electrode

separation, and this property can be expressed in terms of

loss-angle and separation [1], as the product

microradian • millimeters. The effect of electrode surface films on the loss in a 3-terminal gas-dielectric capacitor has been studied for a

variety of surfaces and surface conditions at 1592 Hz, using a state-of-the-art transformer-ratio-arm bridge and

a specially constructed test capacitor. The test capacitor

consists of a parallel-plate guard-ring capacitor whose

electrode separation can be adjusted while maintaining

a vacuum or constant gas condition. The electrode mounting system provides auxiliary adjustments for making the electrodes concentric, coplanar, and parallel. In order to find the product of loss angle and electrode separation of an electrode material, it is necessary that the absolute surface loss of the electrode material for a given

electrode separation be known. This was accomplished by

measuring the losss angle differences between a reference capacitor and the test capacitor at different electrode separations. After applying corrections for the bridge system and

for lead resistance and voltage dependence associated with

the test capacitor, the linear relationship between loss angle

differences and corresponding values of capacitance was

used to calculate the absolute loss angles of the test capacitor

and reference capacitor.

Manuscript received June 15, 1979; revised July 2, 1979.

E. So was a Guest Worker at the National Bureau of Standards, Washington, DC. He is with the National Research Council of Canada, Ottawa,

Ont. K1A OR 8. Canada.

J. Q. Shields is with the Electrical Measurements and Standards Division, National Bureau of Standards, Washington, DC 20234.

Fig. 1. Cross section of test capacitor.

The absolute measurements described were made with

stainless steel electrodes in a dry gas condition. Measure- ments at only one setting of the test capacitor (100 pF) were

subsequently required to calculate the surface characteristics for other electrode materials and relative humidity

conditions.

II. Test Capacitor

A cross section of the test capacitor is shown in Fig. 1 . The

figure is drawn approximately to scale with the guarded

electrode (B) having a diameter of 6.10 cm and the guard

ring (C) having diameters of 6.35 cm and 11.75 cm. The vacuum housing consists of a heavy cylindrical case constructed entirely of stainless steel. The top and bottom of

the housing can be removed to expose the electrode surfaces. Aluminum gaskets are used for making it air tight. Two

windows are provided so that the electrodes can be viewed

from the side and air can be blown through it to clean the

electrodes of dust or lint. The side covers with pipes shown

beside the windows are used to make a closed system for

455
circulating air with a certain relative humidity through the electrode system.

The central guide rod of the movable electrode (A) extends through the top of the housing. The upper end of the guide rod is threaded (10 threads per inch), and a large

diameter knurled nut (spacing wheel) is used on this to set the spacing between the electrodes. This nut works against

the collar of a brass bushing. A bronze pin in the collar fits into a slot in the guide rod and this prevents the rotation of the movable electrode assembly. The rim of the spacing

wheel is divided into 100 slots and a large diameter screw

with fine threads (40 threads per inch) mounted on top of the case (not shown in Fig. 1 ) is used to rotate the spacing wheel

(pushing against one of the slots). The movable electrode (A) is mounted on a metal disk attached to the guide rod by three screws spaced 120° apart. The electrode is insulated from the metal disk and the case by small cylindrical insulators. A metal bellows is attached

to the mounting disk and to the top lid of the case, which fits over the brass bushing.

The guarded electrode (B) and guard-ring (C) are also mounted on a metal disk. The mounting of the guard ring is such that it is insulated from the metal disk at a fixed distance. The guarded electrode is supported by three screws spaced 120° apart. Using these three screws adjustment for coplanarity with the guard ring can be made. The guarded

electrode is held in place by a cylindrical rod at the center of

the electrode, tightened against the metal disk by a spring

and a nut. The hole in the center of the metal disk is made a

bit larger than the rod so that adjustment can be made for eccentricity of the guarded electrode with respect to the guard ring. This whole assembly is again mounted on

another metal disk (see Fig. 1) by three long screws spaced

120° apart, supported by springs. These screws are used for adjusting the parallelism of the upper and lower electrodes. The deviation from perfect alignment which could be

tolerated for each of the mechanical adjustments was deter- mined from theoretical and experimental results. The theoretical calculations were based on earlier work [2] which

showed that conductance resulting from a thin dielectric surface film is proportional to the rate of change of capaci- tance with displacement of the surface. The experimental

determinations consisted of changing the electrode misalignment by measurable amounts while at the same time

increasing the effective dielectric film loss through the use of

vinyl tape applied to the electrode surfaces. The above techniques were also used to show that one half of the loss angle contribution from dielectric surface films is attributable to the unguarded electrode. This result makes

the investigation of different electrode materials much easier since only the unguarded electrode needs to be changed.

III. Absolute Measurement

In order to find the product of loss angle and electrode separation of an electrode material under certain conditions

of measurement, it is necessary that the absolute surface loss of the electrode material for a given electrode separation be

known. The technique used for determining absolute loss is basically the same as that described by Astin and others [3],

[4]. It uses a parallel-plate guard-ring capacitor with variable electrode separation in one arm of the bridge and a standard capacitor in the other arm. The absolute loss angle

of the guard-ring capacitor at different electrode separations

is determined indirectly by first determining the absolute

loss of the standard capacitor. Successive bridge measure- ments are made with the standard capacitor (Cs ) alone or in parallel with auxiliary capacitors (nominal values equal to the standard capacitor Cs and their differences in loss with

respect to Cs known from other measurements) while the capacitance of the guard-ring capacitor is set at Cs , 2CS , 3CS , etc. (in an equal-arm bridge) or 10CS , 20CS , 30CS

, etc. (in a

10: 1 ratio-arm bridge) by changing the separation between

its electrode. The loss angle differences at these settings with

respect to the standard capacitor can then be calculated from the measurements. When the resulting loss angle

differences are plotted against the corresponding values of

the guard-ring capacitance, the points are found to lie on a

straight line. If this line be extended until it intercepts the

axis (guard-ring capacitance = 0), the intercept gives the absolute loss angle of the fixed standard capacitor. This

value can then be used in conjunction with the loss angle

differences to calculate the absolute loss angles of

the guard-ring capacitor.

The linear relationship described above is based on the assumption that the loss angle of the guard-ring capacitor is due entirely to electrode surface losses. In the present work

significant contributions to the loss angle of the guard-ring

capacitor (test capacitor) were found to result from lead

resistances and voltage dependence. These contributions were evaluated and removed from the loss angle differences by means of appropriate corrections. The bridge used to measure loss angle differences is a transformer ratio-arm bridge [5] having a nominal voltage

ratio of 10:1, adjustable over the range ±500 ppm (least count of 1 x 10" 9

) and ±50/irad (least count of 1 nrad)[6].

The test capacitor was used on the low voltage side of the bridge (10 V) and balanced against a 10-pF standard

capacitor on the high voltage side (100 V). Bridge balances were obtained for three different settings of the test capacitor

corresponding to capacitances of 100 pF, 200 pF, and 300

pF, thus requiring two 10-pF auxiliary capacitors which

could be used in parallel with the standard capacitor on the high voltage side. One of the 10-pF fused silica capacitors that maintain the NBS unit of capacitance [7], designated as number 127, was

used as the standard capacitor. Two others, designated as numbers 114 and 109, were used as auxiliary capacitors. If

<5i27> &i i4- a"d (5 109 denote the loss angles of the three fused

silica capacitors and S 100, (5 2 oo> and 6 300 denote the loss angles of the test capacitor at settings corresponding to 100 pF, 200 pF, and 300 pF, respectively, then the bridge balances described previously yield values for the following

loss angle differences: (5 100 - S l21, S l00 - S 114, <5 100 - <5 109, ^200 - (<5i27 + <5io<>)/2, and <5 300 - (<5 127 + <5 114 + <5 109 )/3, from which a simple calculation yields values for the desired

456
2 (LINE)

-.80 H 1 1 1 0 100 200 300 Ctc(pF>

Fig. 3. Difference between the loss angle of the test capacitor and

that of the reference capacitor as a function of the capacitance of

the test capacitor CT <

loss angle differences S l0O -S121, S200 - $121, and

^300 — ^127- The loss angle differences measured by the bridge refer to

loss angles as defined at coaxial tees connected to the line and detector terminals of each capacitor (four-terminal pair

definition [8]). This results from the method used to obtain

loading corrections, i.e., a special extrapolation technique

involving the use of matched cables and connectors [9]. As mentioned previously, corrections are required for loss angle contributions from internal lead resistances and vol- tage dependence. The corrections for internal lead resist- ances are calculated from measurements of each of the

TABLE 1 Estimates of the Effects of Possible Systematic Errors on the Measurement of 6 t21

Sources of Estimate*

Possible Systematic Error (wrad)

1. Loading corrections 0.004

2. Lead resistance corrections 0.005

3. Voltage dependence corrections 0.003

4. Effect of capacitance drift 0.005

5. Bridge calibration and stability 0.005

Direct sum: 0.022

rss: 0.010

*These values are subjective estimates Intended to be comparable with

statistical 2 o limits.

capacitances and resistances shown in Fig. 2. The only

significant distributed capacitance (not shown) was that associated with R2

. One half of this capacitance was

included in C"t2 and an appropriate systematic uncertainty

assigned. The resistances were measured as four-terminal

resistances at 1592 Hz. Small spring contacts inserted through the side windows of the housing provided two

points of contact for each electrode. Corrections for voltage

dependence were calculated from measurements using

previously established standards ofvoltage dependence [10]. The final corrected values for the loss angle differences are

«$ioo 127= -0.580 /irad (1)

<*2oo - <$i27 - -0.385 /irad (2)

^soo -5 127 = -0.192 /irad. (3) The values shown are the averages from eight measurements

involving the use of three dry gases: N2 , C02 , and SF6. There was no statistically significant difference with respect to the choice of gas dielectric. Stainless steel electrodes were

used for all measurements. The linear relationship between

loss angle differences and the corresponding values of capacitance is illustrated in Fig. 3 where the calculated value of the y intercept is

60 -5l21 = -0.774 /irad. (4) Assuming that the values of 6 , 00, 5 200, and <5 300 in (1), (2), and (3) result entirely from surface losses, i.e., assuming that we have successfully identified and corrected for all other

loss angle contributions, then the value of d 0 corresponding

to zero capacitance (infinite electrode separation) is zero and from (4)

6 l21 = 0.774 /irad.

The standard deviation of 5 , 2 7is 0.005 /irad, thus pointing out the fact that individual measurement results deviate

significantly from Fig. 3. Estimates of the effects of possible

systematic errors on the measurement of S l21 are listed in Table I. Items 1 through 3 account for uncertainties in the corrections previously described. Item 4 accounts for uncertainty in the quadrature balance due to capacitance drift (as a result of temperature effects) in association with imperfect

457
adjustment of the phase-sensitive detector. Item 5 is due

mainly to uncertainty in the 10 : 1 voltage ratio of the bridge transformer which was measured using a modified version

[9] of the permutation method [11]. The value obtained for the absolute loss of standard

capacitor 127 was found to agree with the value measured in terms of a toroidal cross capacitor [12] within a few nanoradians. The agreement between measured values provides a valuable check on both absolute measurements. A second

area of agreement is in the value obtained for the surface

characteristic of stainless steel under dry gas condition.

IV. Surface Characteristic Measurements

The surface characteristic (SC) of stainless steel under dry

gas condition can be calculated from the results of the previous section. Substituting the value obtained for <5 127 into (1) yields

<$ 100 = 0.194 /irad.

Multiplying 0.194 /irad by 0.269 mm, the electrode separation of the test capacitor at 100 pF yields

SC - 0.05 /irad • mm.

The surface characteristics of stainless steel under differ- ent relative humidity conditions (to be described shortly) were calculated in the same manner with the only changes

being the numerical values in (1 ). Thus each relative humidity condition requires but one additional bridge balance

with the test capactitor remaining fixed at 100 pF.

The above procedure was also used to determine the surface characteristics of different electrode materials. However, because only the unguarded electrode was

changed, one additional calculation was required, e.g.,

(SC)AA = 2(SC)AS - (SC)*

where (SC)AA is the surface characteristic that would have

been obtained if all electrodes were made of aluminum,

(SC)^ is the value obtained when the unguarded electrode is made of aluminum and the other electrodes are of stainless

steel, and (SC)* is the value obtained under the same

relative humidity condition when all electrodes are made of

stainless steel.

It should be noted that the value of <5 127 does not actually enter into the calculation of surface characteristic, i.e., substituting the expression used to calculate 6 l21 into (1) results in a formula for <5 100 which is independent of This is consistent with the more direct calculation of <5 100 in which (1) is subtracted from (3) and the result divided by 2 yielding

<5ioo = (<Soo - <5ioo)/2 = 0.194 ^rad

which agrees with the value obtained previously.

Thus the one requirement for 3 121 is that its value, whatever it may be, not change during the measurements,

and similarly for the transformer voltage ratio whose value does not enter into the surface characteristic calculations. A

third quantity which must remain constant throughout the measurements is the surface loss of the stainless steel

TABLE n

Surface Characteristics of Different Electrode Materials Under

Dry Gas Condition

Electrode Material Surface Characteristic

(urad-mm) Stainless steel 0.05

Silver-plated brass 0.06

Gold-plated brass 0.06

Nickel -plated brass 0.06

Rhodium-plated brass 0.08

Brass 0.12

Aluminum 0.10

reference electrodes consisting of the guarded electrode and

guard ring. An estimate of the overall stability of these

quantities was obtained by periodically remeasuring the

original system, i.e., all electrodes of stainless steel and under

dry gas condition. As previously mentioned, surface characteristics were measured over a wide range of humidity. A convenient

method using a saturated salt solution exists for establishing

atmospheres of known relative humidity which depend

upon the equilibrium vapor pressure of water when a chemical is dissolved in it [13], [14]. A closed circulating air system incorporating a suction-pressure pump was used in conjunction with five different saturated salt solutions to obtain relative humidities of 14, 33, 54, 75, and 92 percent. Table II shows the surface characteristics of different electrode materials under dry gas (N 2 ) conditions. Fig. 4 shows the surface characteristics of stainless steel, brass, and

of silver, gold, nickel and rhodium-plated brass under

different relative humidity conditions. A semilog scale is used in Fig. 5 to show the surface characteristics of alu- minum and of a particular specimen of anodized aluminum

under the same relative humidity conditions; stainless steel

is included for comparison purposes.

Possible systematic errors in the surface characteristic determinations are the same as those listed in Table I but with each having a numerical estimate of 0.002 /irad mm or

less. An overall uncertainty of 0.01 /irad mm was assigned

to the surface characteristic measurements based on the esti- mates of possible systematic error (root-sum-square =

0.004 /irad mm) and a standard deviation of 0.003

/irad mm. It is important to realize that these uncertainties apply only to the measurement process and that the quantities being measured are dependent upon surface conditions which can be difficult to control or specify.

It should be noted that cleaning the electrodes and

keeping them clean was quite necessary for consistent

results. Small quantities of dust were especially troublesome. The electrodes were prepared by first degreasingthem using acetone and methanol, then washing them in soapy water

(using a detergent) and rinsing them with distilled water. To

avoid water spots methanol was again used and dry air was

then blown over the surface of the electrodes.

458
• 286

I I 1 1 I I I I I I I

I 0 20 30 40 50 60 70 80 90 100

R. H. (%)

Fig 4. Surface characteristics (SC) of stainless steel, brass, and of silver, gold, nickel, and rhodium-plated brass as a function of relative humidity

(RH).

V. Conclusion

Stainless steel (type 304) seems to have the best surface

characteristic as compared to the other electrode materials under dry conditions as well as under moisturized conditions. The surface characteristic of stainless steel under dry

lOOCOOr

0 01 1 1 1 I i I I l__i i i 10 20 30 40 50 60 70 60 90 100 R H (% )

Fig. 5. Surface characteristics of aluminum, anodized aluminum, and

stainless steel as a function of relative humidity (semilog scale).

gas condition is 0.05 /^rad • mm, while the surface characteristics of the silver, gold and nickel-plated brass are only

slightly higher at 0.06 nvad • mm. Thus under dry conditions these four materials are recommended for use as electrodes in low-loss reference standards. Even though

aluminum under dry conditions is slightly better than

unplated brass, aluminum would be a questionable choice as the electrode material of a capacitor which is to serve as a

low-loss reference standard. In the presence of moisture the

surface losses of aluminum increase much more rapidly than

for the other measured electrode materials. In addition, it was found that the surface of aluminum releases absorbed

moisture less readily than the surfaces of the other materials

studied.

References

[1] A. V. Astin, "Nature of energy losses in air capacitors at low frequencies," J. Res. Nat Bur. Stand., vol. 22, p. 673, June 1939, RP1212.

[2] J. Q. Shields, "Phase-angle characteristics of cross capacitors," IEEE

Trans Instrum. Meas., vol. IM-21, pp. 365-368, Nov. 1972.

[3] A. V. Astin, "Measurement of relative and true power factors of air capacitors," J Res. Nat. Bur. Stand., vol. 21, p. 425, Oct. 1938, RP1138.

[4] W. B. Kouwenhoven and E. L. Lotz, "Absolute power factor of air capacitors," AlEE Trans., vol. 57, p. 766, 1938.

[5] A. M Thompson, "The precise measurement of small capacitances," IRE Trans. Instrum., vol. 1-7, pp. 245-253, Dec. 1958.

[6] R D. Cutkosky, "Active and passive direct-reading ratio sets for the comparison of audio-frequency admittances," J. Res. Nat. Bur.

Stand., vol. 68-C, no. 4, pp 227 236, Oct. -Dec. 1964.

459
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT. VOL. IM-28. NO. 4. DECEMBER 1979

[7] R. D Cutkosky and L. H. Lee, "Improved (en-picofarad fused silica dielectric capacitor." J. Res. Nat. Bur. Stand., vol. 69-C, pp. 173-179.

July-Sept. 1965.

[8] R D. Cutkosky. "Four-terminal pair networks as precision admittance and impedance standards." Commun. Electron., vol. 70, pp. 19-

22, Jan. 1964.

[9] J. Q Shields. "Measurement of four-pair admittances with two-pair

bridges." IEEE Trans. Instrum. Meas.. vol. IM-23. pp. 345-352, Dec.

1974.

[10] , "Voltage dependence of precision air capacitors." J. Res. Nat. Bur. Stand., vol. 69-C, no. 4, pp. 265-274. Oct.-Dec. 1965.

[11] R D Cutkosky and J. Q. Shields. "The precision measurement of transformer ratios," IRE Trans. Instrum.. vol. 1-9. pp. 243-250. Sept. 1960.

[12] J. Q. Shields. "Absolute measurement of loss angle using a toroidal cross capacitor," IEEE Trans. Instrum. Meas.. vol. IM-27, pp. 464-466. Dec. 1978.

[13] A. Wexler and S Hasegawa. "Relative humidity—temperature rela- tionships of some saturated salt solutions in the temperature range 0° to 50°C," J. Res. Nat. Bur. Stand., vol. 53. no. 1. July 1954, RP2512.

[14] A. Wexler and W. G. Brombacher. "Methods of measuring humidity and testing hygrometers." NBS Circular S12, Sept. 28. 1951.

460
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-27, NO. 4, DECEMBER 1978 A Programmable Phase-Sensitive Detector

for Automatic Bridge Applications

ROBERT D. CUTKOSKY, fellow, ieee

Abstract—A phase-sensitive detector is described which features programmable gain, digital output, and a resolution of 0.02 uV. The

instrument suppresses harmonic sensitivity through the use of analog

circuitry which multiplies the signal by constant amplitude sine and

cosine waves derived from an external reference of arbitrary voltage and frequency.

I. Introduction

ANAUTOMATIC audio frequency bridge is now under

construction at the National Bureau of Standards

(NBS). Although this instrument is intended mainly for calibrating capacitors, ac resistors, and inductive dividers in conjunction with the NBS calibration service, it also serves

as a convenient vehicle for investigating techniques for achieving rapid convergence of bridges containing multiple

auxiliary balance circuits. Conceptually the bridge is balanced by measuring the complex voltages at a number of points with a phase

sensitive detector or voltmeter, and by adjusting the main

and auxiliary bridge variables until all of these voltages are

nulled, or are sufficiently small. This process may be iterative, and is performed under computer control. Many commercial phase sensitive detectors are available

for use with manually balanced bridges, but none were

found which combined programmable gajn and digital

output with either very low intrinsic harmonic sensitivity or a means of automatically tuning the signal channel to suppress harmonic sensitivity. The instrument described here was designed to meet these requirements.

II. Design

This instrument was intended for use with a nominally

frequency-independent bridge driven by a frequency synthesizer with a reasonably pure sine-wave output. The

harmonic rejection requirements of such a system are not

especially severe, but the use of switching type demodulators, which are highly sensitive to odd harmonics, would

have required the use of a tuned signal amplifier. This complication was avoided by using analog multipliers for the in-phase and quadrature demodulators (see Fig. 1). The

signal amplifier is untuned, and connected to the x inputs of both multipliers. The y input of the first multiplier is driven with a sine wave having a peak voltage of 5 V derived from

Manuscript received June 5, 1978. The author is with the Electrical Measurements and Standards Division, National Bureau of Standards, Washington, DC 20234.

SIGNAl INPUT PRE MAIN MAIh

INPUT ATTEN AMP ATTEN AMP

r-vw-,

1

REF AMP AGC ABS TV> INTEG THRESHOLD

INPUT MULT VALUE 1 DETECTORS

BIAS

Fig. 1. Block diagram of phase sensitive detector.

the reference oscillator; the y input of the second multiplier

is driven with a cosine wave having a peak voltage of 5 V

derived from the same oscillator. The output of each multiplier is integrated for one second and these averaged values

are converted to digital form upon reception of a local or remote conversion command with a pair of 8-bit analog to

digital converters. Use is thereby made of the orthogonality

of the sine and cosine functions to suppress harmonic

sensitivity. The integration time was chosen to be one second in our

instrument in order to achieve an adequate signal to noise

ratio. A capability for selecting a shorter integration time when less than maximum sensitivity was required would

have resulted in a generally faster instrument, but in addition to the increased complexity entailed, an integration

time shorter than one second would have caused averaging

uncertainties at low frequencies.

The constant amplitude sine wave is obtained through the use of an automatic gain control circuit in which the gain

control element is an analog multiplier. The difference between a dc reference voltage and the absolute value of the output sine wave is integrated and applied to the y input of the multiplier for level control. In order to accommodate

reference input voltages from 0.06 V to 40 V, one of five discrete ranges is automatically selected dependent upon the

states of upper and lower threshold detectors fed by the

integrator.

The constant amplitude cosine wave is obtained by

integrating the constant amplitude sine wave and using the same gain control circuitry described above, except that the

U.S. Government work not protected by U.S. copyright

461
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-27, NO. 4, DECEMBER 1978

threshold detectors select different integrating capacitors to

yield a useful frequency range of 10 Hz to 50 kHz.

If one of the thresholds is crossed, the appropriate status is displayed, and signaled to the computer if in "remote" mode.

Then, an internal one-second clock shifts resistors or capacitors into or out of the appropriate circuit as required, until both integrators are within their prescribed limits. After a

further delay to provide adequate settling, the status indication is cleared. The signal amplifier consists of a preamplifier containing a low-noise field effect transistor [1], a programmable

attenuator with a range of 75 dB and a post amplifier. An

additional 60 dB of attenuation can be obtained by swit- ching a 1000:1 divider into the circuit before the

preamplifier. A TTL decoder is provided so that the attenuation can be programmed in standard binary code, the least significant bit representing 5 dB. Maximum gain is 135 dB,

which gives a resolution of0.02 /iV. At minimum gain (OdB),

full-scale indication represents about 10 V. The instrument is housed in a 5i" relay rack module,

which includes a double mu-metal shield for the

preamplifier, the necessary power supplies, the- logic circuits to provide timing, decoding and status information, and the LED indicators for local display of gain, output, and status.

References

[1] S. Harkness, "Low noise FET amplifiers for low loss capacitative sources," NPL Report DES 41, National Physical Laboratory, Teddington, England, June 1977.

462
Reprinted by permission from IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT

Vol. IM-23, No. 4, December 1974, pp. 345-352 Copyright 1975 by the Institute of Electrical and Electronics Engineers, Inc. PRINTED IN THE U.S.A.

Measurement of Four-Pair Admittances with Two-Pair Bridges

JOHN Q. SHIELDS

Abstract—The purpose of this paper is twofold: 1) to show how

extrapolation techniques can be used to obtain well-defined and

versatile measuring systems; and 2) to apply such techniques to the measurement of four-pair admittances with two-pair bridges. When

four-pair admittances are large, measurements with two-pair bridges have limited precision, but with reasonably small admittances, measurement precision is essentially equal to that attained with

four-pair bridges.

INTRODUCTION

ALTHOUGH a major portion of the present paper will be devoted to the measurement of four-pair ad- mittances with two-pair bridges, this represents but one example of the use of extrapolation techniques which are applicable to a variety of situations. Basically, extrapolation techniques provide a means for adapting existing

measuring systems to changing needs while at the same

time, retaining well-defined measuring conditions. The

paper illustrates this, to a degree, by including applications which are not restricted to four-pair admittances,

e.g., the elimination of the effects of matched cables, and

a double permutation method for measuring transformer

ratios. When the applications are restricted to four-pair

admittances, it is important to keep in mind that any

two-pair admittance is easily converted to a four-pair

admittance by the addition of two coaxial tees. The reader

Manuscript received July 3, 1974; revised August 22, 1974. The author is with the Electricity Division, National Bureau of Standards, Washington, D. C. 20234.

is referred to earlier papers [1], [2] by Cutkosky for a

discussion of the advantages of four-pair admittances as

precision standards. When four-pair admittances are large, measurements

with two-pair bridges have limited precision, but with

reasonably small admittances, measurement precision is essentially equal to that attained with four-pair bridges. For example, measurements of four-pair capacitance standards in the range from 10 to 1000 pF at « = 104 can be made with uncertainties on the order of 1 or 2 parts in 109 using either four-pair bridges or two-pair bridges.

The author has attempted to write the paper in a manner

that would be useful to practitioners of precision measure- ments at many levels. As a result, there is a detailed description of the direct-reading ratio set and two-pair bridges

at one extreme, and, at the other extreme, the term

"defining transformers," 1 is used here with only a refer- ence for descriptive purposes. The reader's discretion is encouraged.

THEORY

The four-pair admittance, as originally described by

Cutkosky [1], [2], is shown in Fig. 1. The parameter of

interest is defined as the ratio of the current at pair 3 to

1 The auxiliary balances of the four-pair DRRS are such that four- pair admittances are defined at "defining transformers" [2] rather than at connectors. This change in defining conditions was easily accounted for by adding 2.35 pF to the measured values of the twoterminal admittances which were extrapolated to zero.

463
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1974

Fig. 1. Complete circuit of four-pair admittance standard.

the open-circuit voltage at pair 2, subject to the condition

that both the voltage and current at pair 4 are zero. This

parameter has the dimensions of admittance and may be

written

P„J4 — * 32 —

e2 H = it = 0, e4 = 0.

Using Fig. 1, it can be shown that P32

14 = Fr (l + F2 Z2 )(1 + Y3Zs).

(1)

(2)

In comparison, the corresponding two-pair admittance

is defined as the ratio of the short-circuit current at pair 3 to the voltage at pair 2, with the implicit condition that

pairs 1 and 4 are open-circuited (capped if necessary- otherwise ignored). This may be written

F32 =

e2 e3 = 0. (3)

If admittances are added to pairs 1 and 4, the value of F3 > will be changed, but Pwu will remain unchanged. This

applies even if the terminals of pairs 1 and 4 are modified

in the process.

It can be shown that if the admittances added to pairs 1 and 4 are correctly chosen, then

F32 = /V4

. The correct admittances are those which will result in F22 = F33 = 0 where

F22 =

and

F33 —

e3

e3 = 0

ei = 0.

(4)

(5)

F22 is the easily measured two-terminal admittance at

pair 2 with pair 3 shorted. For most purposes, F33 can be

measured in a similar manner, i.e., as the two-terminal

admittance at pair 3 with pair 2 shorted. This changes

the condition e\ = 0 in (.">) to e2 = 0 and results in F32 = P32

14 + terms in ( YZ) - and higher. (6)

It is not generally practical to adjust F22 = F33 = 0, but fortunately, it is sufficient to add an admittance to

pair 1, note the corresponding changes in F22 and F32 , and effectively extrapolate F22 to zero by means of appropriate corrections to F32 . A corresponding sequence

is used to effectively extrapolate F33 to zero.

Fig. 2. Comparison of two-pair admittances using the two-pair

direct-reading ratio set.

APPLICATIONS

Although the basic concepts presented in the previous

section should be applicable to a variety of measuring

systems, the examples presented in this paper have been

limited to measurements using the transformer ratio-arm

bridge [3], [4] with adjustable ratio. This type of bridge

is commonly termed an ac direct-reading ratio set [5]

and will henceforth be referred to simply as a DRRS.

In Fig. 2, the equivalent circuit of a DRRS is represented

by the component having coaxial connectors .41 and .42. The bridge is balanced by adjusting the voltage ratio at connectors .41 and .42 by means of a small adjustable

voltage designated e in Fig. 2. The voltage e consists of two components, a and /J, which provide adjustments to both the real and quadrature components of the voltage

ratio. Typically, a and /3 would each consist of six decades

providing a range of ±/>00 ppm and ±f>00 /arad, respectively, and a least count of 1 X 10-9 and 1 nrad, respectively.

The complete circuit of Fig. 2 is typical of the type of circuit used to compare precision two-pair admittances

whose nominal values are in a ratio of 10:1. Net currents

in the cables are reduced to negligible amounts by the use of coaxial chokes [3], [(>], which are designated Ch

in Fig. 2. The component having connectors B2 and B?> represents the equivalent circuit of a two-pair admittance

defined at connectors B2 and 53, and similarly for the component having connectors C2 and C3. It will be as- sumed for the present that the coaxial tee is ideal, i.e., the voltage at Dl and D2 is zero when the voltage at Z)3

is zero.

A. Four-Pair Measurements

Fig. 3 is identical to Fig. 2 if all adjacent connectors

are joined. If Fl and F2 are disconnected, the component

having connectors Bl, B2, B'.i, and BA can be considered

as a pictorial representation of a four-pair admittance

(henceforth termed four-pair admittance Pb) whose complete circuit would be similar to that shown in Fig. 1 with pairs 1, 2, 3, and 4 corresponding to connectors Bl, B2, B3, and B4, respectively. Similar considerations can be applied to the component having connectors CI,

C2, C3, and C4 (henceforth termed four-pair admittance Pc )

464
shields: four-pair admittances

Fig. 3. Comparison of two-pair admittances having adjustable shunt admittances F2 , F3 , F6 , and F6 .

It is important to bear in mind that when all connectors are joined, Fi, F2 , and F3 represent the lump-sum

admittances of the equivalent circuit of a two-pair admittance, and, similarly, for F4 , F5 , and F6 . If F2 , Y3 , F5 , and F6 have been adjusted so that F2 = F3 = — Fi and

F5 = F6 = — F4 , with all connectors joined, then, refer- ring to (4) , (5) , and (6) , it can be seen that the value of e required to balance the bridge will yield a measure of the

ratio of Y\ to F4 which is the same2 as the ratio of fourpair admittance PB to four-pair admittance Pc , as would

be obtained with a regular four-pair bridge. Xote that when the bridge is balanced, there is no current between

-41 and B2 or between A2 and C2. This not only satisfies one of the defining conditions of a four-pair admittance,

but also eliminates the need to account for the equivalent

lead impedances in the DRRS.

As pointed out in the section on theory, it is not neces- sary to reduce the current between A 1 and B2 to zero.

It is sufficient to change the current, note the corresponding change in e required to rebalance the bridge, and

effectively extrapolate the current to zero by means of appropriate corrections to e. Details of the complete

extrapolation are more easily described using Fig. 4 and

the equation

e* = e.v + {-Ym/YL ){ein - eN ) + ( - Ym/YL) (e„t - eN ) + ( — Fes/Fi,) (eci - eN ) + (-YC3/YL ) [eCi - eK)

(7)

where ex is the value of e which yields a measure of the

ratio of four-pair admittance P« to four-pair admittance

Pc , e,v is the value of e required to balance the bridge in the normal condition shown in Fig. 4 (unused connectors may require caps which are not shown), YL is a twoterminal admittance defined at a connector which is joined to connectors Bl, B\, CI, and C4 (each in turn),

e m, eBi , ecu and ecu are the corresponding values of e required to balance the bridge, Ym, Y„3 , FC2, and FC3 are two-terminal admittances measured at connectors B2, B3,

2 Neglecting the effect of terms in (YZ)2 and higher. The remainder

of the paper is subject to the preceding condition wherever applicable.

Fig. t. Four-pair admittances connected into a two-pair bridge.

C2, and C3, respectively, with a short at connectors B'.i, B2, C3, and C2, respectively. For practical usage, (7) must be separated into real and imaginary parts yielding two equations, one for ax and one for where a and 0 correspond to the real and

quadrature components of e as previously described. However, for simplicity, (7) will be used in the discussion which follows. Note that the last four terms of (7) are simply corrections which are applied to eN . In general, the corrections do not need to be determined to high accuracy, e.g., ±1 X 10~3 is sufficient for many applications. If YL is chosen to be many times larger than Ym, Ym, Yc2, and

FC3, then the measurements of (e M — e^) and (e C\ — eN ) will be above the noise level of the system; the effect on

the measurements of (em — eN ) and (e C \ — ca>) will de- pend on the particular circuit being used. YL should

obviously not be a component which reduces the signal- to-noise ratio unnecessarily.

B. Two- and Three-Pair Measurements

If the third and fourth correction terms in (7) are eliminated, ez will yield a measure of a two-pair admittance

in terms of a four-pair admittance. If the second and

fourth correction terms in (7) are eliminated, ex will yield a measure of the ratio of two three-pair adm ttances. The

latter is particularly useful for comparing the main ratios

of different transformer ratio-arm bridges (including fourpair bridges which are adaptable to a three-pair configuration).

C. Elimination of Cable Effects In all of the examples thus far presented, the admittances have been defined at the ends of cables connected

to the admittance standards. This can cause problems

under certain circumstances, especially since the cables containing coaxial chokes are of necessity quite long

(often approaching 3 m in length).

Fig. ") is intended to illustrate two four-pair admittance

standards, Pu and Pc, whose defining pairs are coaxial connectors (Bl, B2, B'.i, B4, and CI, C2, C3, C4) which

are permanently attached to the housings of the standards. The component having connectors (71, 02, and G3 rep- resents a cable with a permanently attached coaxial tee (henceforth termed cable G) and similarly for the component having connectors HI, 7/2, and HA (henceforth

465
IKEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1974

Fig. 5. Circuit for illlustrating how to eliminate the effects of coaxial cables and coaxial tees used to connect the admittance standards

to the bridge.

termed cable H) . The other cables will be termed cable I and cable J. The coaxial tee (Dl, D2, D3) will no longer

be assumed to be ideal. It can be shown that the effects of cables G, H, I, and J, coaxial tee (Dl, D2, D3), and the equivalent lead impedances of the DRRS can all be eliminated by extrapolation techniques. Details of the extrapolation are more

easily described using the equation

ex = eN + {—Ym/YL)(eDi — eN) + ( — YBz/YL )(eBi — eN ) + (-YC2/YL)(eci - eN) + (-YC3/YL)(ec< - eN) + (-Y0i/YL)(eG3 - eN ) + (-YHi/YL)(em -eN ).

(8) Note that with the exception of the two extra terms at the end of (8), it is identical to (7). These two terms are

correction terms similar to the four correction terms which

precede them, with egi and e#3 being the values of e required to balance the bridge when the defining connector

of YL is joined to connectors G3 and H3, respectively.

Ygi and Yh\ are the two-terminal admittances measured

at connectors Gl and HI, respectively, with connectors G2 and H2 open-circuited.

One result of applying the correction terms is that, in

effect, no current flows between Al and Gl, between G2

and 52, between A 2 and HI, or between H2 and C2. This means that only the lead impedances and shunt

admittances associated with cables G and H affect the changes in voltage from Al to G2 and from A2 to H2,

and these changes are proportional to the voltages. If cables G and H are matched cables, i.e., if they have

identical electrical characteristics, then the ratio of the

voltage at G2 to the voltage at H2 will be the same as the

ratio of the voltage at A 1 to the voltage at A2.

A similar effect occurs at the detector end of the circuit, only this is related to proportional changes in current

rather than voltage. The result is that if cables / and J

are matched cables, and if the coaxial tee is such that Dl

and D2 are electrically symetrical with respect to D3, then

they will have no effect on the bridge balance. It is in- teresting to note that matched cables yield equal lead impedances, whereas with ordinary unshielded bridges

having 10 to 1 ratio arms, the lead impedances would

need to be in the ratio of 10 to 1 with respect to each other

if they were to have no effect on the bridge balance. If the

cables are not matched, and if the coaxial tee is not

symmetrical, then we can interchange cables G and H,

cables / and J, and rotate the coaxial tee. This will result

in a new value of ex . It can be shown that the average of the two values of ex will yield a correct value for the ratio of the two four-pair admittances.

D. Cable Effects in Other Circuits The extrapolation techniques for eliminating the effects of cables G and H are not limited to the circuit shown.

For example, if the two four-pair admittances are con- nected into a regular four-pair bridge, we can still add

admittance YL to G3 and to HZ, note the corresponding

changes in the bridge balance, and apply correction terms

similar to those described previously.

If the components containing Yl and F4 represent

two-pair admittances, then the effects of cables G and H

can only be partially eliminated. In general, however, if the junctions of G3 and H3 are close to G2 and H2, respectively, then the effects not eliminated will be quite small compared to those eliminated, which include the effects of the equivalent series irnpedances of the DRRS. When

Yl is added to G3, the acfrnittance to be extrapolated to zero is the sum of YGi and YUi (as previously defined)

and similarly for the other side of the bridge. If the

effects of cables / and / are large, coaxial tees can be

joined to II and Jl, thus permitting an extrapolation

process similar to the preceding.

E. Parallel Admittance Measurements

Fig. 6 shows two nominally-equal four-pair admittances

in parallel being compared to a third four-pair admittance.

Although the auxiliary balances associated with regular

four-pair bridges are usually not adaptable to this type of comparison, the comparison by means of extrapolation

techniques is the same as described previously, with the

six resulting correction terms similar to the four correction terms in (7). If matched cables are used (not shown in

Fig. 6) , the extrapolation process results in nine correction terms which are similar to the six correction terms in (8)

If the cables are not perfectly matched and if the coaxial

star {Ml, M2, M3, Mi) is not perfectly symmetrical, then

the cables and the star can be permuted to obtain a total

of three different values of ez . It can be shown that the average of the three values will yield a correct measure of the sum of two four-pair admittances in terms of a third

four-pair admittance.

In order to better understand the permutation process, imagine that three cables similar to G in Fig. 5 are inserted between A 1 and B2, between A 2 and K2, and between A 3 and L2, and that each cable is used to join a different pair

during each of the three different measurements. The other

half of the extrapolation process, which is carried out

simultaneously, is more easily understood by imagining

that Ml, M2, and 71/3 are connectors at the ends of three

cables which have been permanently joined to a coaxial

star, and that each of these connectors is joined to a

different four-pair admittance during each of the three

466
SHItLDS: FOUR-PAIR ADMITTANCKS

Fig. 6. Circuit for measuring the sum of two four-pair admittances

in terms of a third four-pair admittance.

measurements. It is assumed that any difference in the open- circuit voltages at connectors A2 and ^43 has been

accounted for. The techniques just described are not limited to the

circuit shown in Fig. 6. These techniques can be extended

to circuits where the two four-pair admittances containing

F7 and F8 in Fig. 6 are replaced by three or more four-pair

admittances in parallel. Furthermore, it is not necessary

that all of the four-pair admittances in parallel be nominally equal to each other. The most important application

of this is where one of the four-pair admittances is a small

adjustable admittance which is used to balance the bridge.

Thus the DRRS used in the examples throughout this paper can be replaced by a regular bridge transformer having a fixed voltage ratio.

F. Transformer Ratio Measurements

Fig. 7 shows part of the circuit used at NBS for measuring the nominally 10 to 1 ratio of precision bridge trans- formers. The remainder of the circuit consists essentially

of the bridge transformer whose ratio is to be measured,

plus eleven nominally-equal capacitors. The method to be

used is a modification of the permutation method which was described in an earlier paper [7], and the reader is referred to this paper for further details. Fig. 7(a) and (b) shows two different views of the same setup. The component in the lower right-hand

corner of Fig. 7(b), having eleven toggle switches, is termed a load adjuster. The load adjuster consists of eleven metal boxes, each having two coaxial connectors, one on the top and one on the bottom. In practice, the

load adjuster is mounted on the edge of a table so that

cables can be attached to the bottom coaxial connectors. Each metal box contains a trimmer capacitor and a

relatively large capacitor which are, in effect, connected

between the inner and outer conductors of the cables. Each of the large capacitors, which correspond to Yl in the previous discussion, can be removed from the circuit by means of the appropriate toggle switch. Each of the eleven coaxial connectors on the bottom of the load adjuster is joined to a coaxial cable (not shown

in Fig. 7) which contains a coaxial choke and is joined to one of eleven 10-pF fused silica capacitors. The eleven

detector cables from the capacitors are joined together

with a twelfth cable, which is connected to a phase sensitive detector. There are no special requirements for joining the detector cables [7]. The eleven cables from the top of the load adjuster are plugged into eleven of the twenty-two double-ended

coaxial connectors, which, together with the bakelite board on which they are rigidly mounted, is termed a

plug board. Looking at the other side of the plug board

(Fig. 7(a)), note that the cables on the top row are joined to eleven coaxial connectors symmetrically placed near the outside edge of one of the hollow metal disks. The cables on the bottom row are joined to eleven connectors similarly placed on the other disk. Each disk has two additional coaxial connectors which are centrally

located, one on the top and one on the bottom. The

center conductors of all thirteen coaxial connectors are soldered to a thin metal disk located inside the hollow

disk. The resulting component consists oi thirteen coaxial connectors whose inner conductors are joined together

by means of a solid metal disk and whose outer conductors

are joined together by means of a hollow metal disk. This

will henceforth be referred to as a coaxial star. Note that the four darker colored cables are joined to the centrally-located coaxial connectors of the coaxial

star. In practice, the connectors at the ends of the two

shorter cables are rigidly mounted to some solid object

so as not to disturb the other cables, and it is at these connectors that additional two-terminal admittances are added, i.e., similar to Yl in the previous discussion. The

connectors at the ends of the two longer cables are joined

to the bridge transformer DRRS whose ratio is to be

measured. For illustrative purposes, we will assume that the connectors are joined to connectors A 1 and A2 of the DRRS,

which has been used in the figures throughout this paper. Measuring the ratio of the DRRS actually amounts to finding ea , where ea is the value of e which will result in a perfect 10 to 1 open-circuit voltage ratio at connectors Al and A2.

The basic permutation method consists of eleven bridge balances where each capacitor, in turn, is connected to the

high-voltage side of a 10 to 1 transformer, and the other

ten capacitors are connected to the low-voltage side. Note

that in Fig. 7(b), capacitors 1-10 are plugged into connectors 1-10 of the bottom row of the plug board, and

capacitor eleven is plugged into connector eleven of the top row of the plug board. This corresponds to one of the

eleven bridge balances referred to previously. In the second of the eleven balances, capacitors 1-9 are plugged

into connectors 1-9 of the bottom row of the plug board,

capacitor ten is plugged into connector ten of the top row

of the plug board, and capacitor eleven is plugged into connector eleven of the bottom row of the plug board. Proceeding in the same manner, we obtain eleven values

of e required to balance the bridge in each of the eleven

configurations.

It can be shown that the average of the eleven values

of e is equal to e0 , provided that lead impedances and the

467
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1974

equivalent series impedances of the DRRS are negligible

and the voltage dependence of the capacitors has been

accounted for [8]. If the impedances are not negligible, we can obtain the same result by using matched pairs of coaxial star/cable combinations, while, at the same time,

extrapolating the current to zero at connectors A 1 and

A2 of the DRRS and also at the capacitor end of each of the plug-board connectors. The star/cable combinations

include all of the circuit shown in Fig. 7, with the exception

of the load adjuster and the eleven cables connected to it. If the star/cable combinations are not matched, they

then can be interchanged, the measuring procedure re- peated, and a second value of e0 determined. It can be

shown that the average of the two values of e0 will yield a correct value of e0 . Interchanging the star/cable combinations amounts to interchanging the connectors joined to Al and A2 of the DRRS and, of necessity, changing the plug-board connections so that ten capacitors are plugged

into the top row of connectors and one capacitor is plugged

into the bottom row of connectors. In practice, the two star/cable combinations were

found to be better matched than had been anticipated.

The two values of e„ differed by less than 1 part in 109

. More specifically, the two values of a„ (the real component

of e0 ) differed by less than 1 part in 109 and the two

values of 0O (the quadrature component of e0 ) differed by

less than 1 nrad. In addition to the star/cable combinations being well matched, the design and construction of the stars was such

that the effects of mutual impedances associated with the

extra currents in the low-voltage star were both small and

uniform. As a result, it was found that the load adjuster could be eliminated from this particular circuit since it was not necessary to extrapolate the currents to zero at the plug- board connectors, nor was it necessary to use the trimmer capacitors in the load adjuster. Stating the same

thing in a slightly different manner, the extrapolation of currents to zero at the plug-board connectors results in a correction to a0 of 2 parts in 1010 and a correction to 0O of 1 X 10-10 rad, and, since the corrections result from

mutual impedances in the stars, the corrections are applicable with or without the use of the load adjuster.

468
shields: four-pair admittances

Fig. 7. Part of the circuit used to measure the 10 to 1 ratio of a

bridge transformer or DRRS (two-pair or four-pair). Two views

of the same setup are shown.

The circuit of Fig. 7 was also used to measure the 10 to 1 ratio of a four-pair DRRS [2] whose current cables were joined to the centrally-located connectors on the top

of the stars and whose potential leads were joined to the bottom connectors of the stars. The only corrections required were for mutual impedances as described previously.

G. Comparison of Measuring Systems

As a check on extrapolation techniques in general,

four-pair capacitance standards in the range from 10-

1000 pF were compared at co = 104 using both the four- pair DRRS2 and the two-pair DRRS. Maximum discre- pancies between the two types of comparisons were 2 parts

in 109 for the capacitance and 2 nrad for the phase angle.

All discrepancies could be accounted for by sensitivity and

stability of the various components.

CONCLUSION

The use of extrapolation techniques to provide well- defined and versatile measuring systems has been illus- trated by means of a number of applications. Note that

bridge transformer ratios are measured (application F)

in the same manner in which they will later be used to measure admittance ratios (applications A and B). The

admittance ratios so determined can, in turn, be used to measure the main ratios of other bridges, or of the same

bridge, when its defining connectors have been changed

by the addition of cables and/or stars. A change in the

value of an admittance standard resulting from a change

469
IEEE TRANSACTIONS ON INSTRUMF.NTATION AND MEASUREMENT, VOL. IM-23, NO. 4, DECEMBER 1974

in its denning connectors can be measured in a similar manner provided that the admittances to be extrapolated

to zero are properly accounted for. Note also that ' 'application E" provides a second method

for measuring precision transformer ratios. This method

should be of special interest when precision transformers

are available, but large numbers of nominally equal

admittances, as required in "application F," are not

available.

It is hoped that four-pair admittanced will have a more

practical appeal and receive wider usage now that they can be measured with two-pair bridges (including commercial bridges). At the other extreme, as pointed out by

Cutkosky, combining extrapolation techniques with four- pair bridge techniques provides exciting possibilities for data acquisition and automation at the highest levels of

precision.

REFERENCES

[1] R. D. Cutkosky, "Four-terminal pair networks as precision admittance and impedance standards," Commun. Electron., vol. 70, pp. 19-22, Jan. 1964.

[2] , "Techniques for comparing four-terminal pair admittance

standards," J. Res. Nat. Bur. Stand., vol. 74C (Eng. and Instr.), No. 3 and 4, pp. 63-78, July-Dec. 1970.

[3] A. M. Thompson, "The precise measurement of small capaci- tances," IRE Trans. Instrum., vol. 1-7, pp. 245-253, Dec. 1958.

[4] M. C. McGregor et al, "New apparatus at the National Bureau

of Standards for absolute capacitance measurement," IRE Trans.

Instrum., vol. 1-7, pp. 253-261, Dec. 1958.

[5] R. D. Cutkosky, "Active and passive direct-reading ratio sets

for the comparison of audio-frequency admittances," J. Res. Nat.

Bur. Stand., vol. 68C (Engr. and Instr.), no. 4, Oct.-Dec. 1964.

[61 D. N. Homan, "Applications of coaxial chokes to ac bridge cir- cuits," J. Res. Nat. Bur. Stand., vol. 72C (Eng. and Instr.), no. 2, pp. 161-165, Apr.-June 1968.

[7] R. D. Cutkosky and J. Q. Shields, "The precision measurement of transformer ratios," IRE Trans. Instrum., vol. 1-9, pp. 243-250,

Sept. 1960.

[8] J. Q. Shields, "Voltage dependence of precision air capacitors,"

J. Res. Nat. Bur. Stand., vol. 69C (Eng. and Instr.), no. 4, pp. 265-274, Oct.-Dec. 1965.

470
Reprinted by permission from IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT

Vol. IM-21, No. 4, November 1972, pp. 365-368 Copyright 1972, by the Institute of Electrical and Electronics Engineers, Inc. PRINTED IN THE U.S.A.

Phase-Angle Characteristics of Cross Capacitors

JOHN Q. SHIELDS

Abstract-It has been found that under proper conditions thin di- electric films on the electrodes of certain types of cylindrical cross capacitors tend only to produce equal and opposite contributions to the phase angles of the two cross capacitances. To a lesser degree, this

Manuscript received June 26. This work was supported in part by the National Bureau of Standards. The author is with the Absolute Electrical Measurements Section,

Electricity Division, Institute for Basic Standards, National Bureau of

Standards, Washington, D.C. 20234.

same type of cancellation effect has been found to be associated with

toroidal cross capacitors.

I. Introduction

NORMALLY, the greatest uncertainties in a determination of the true phase angle of a three-terminal air ca- pacitor result from the effects of thin dielectric films on the electrode surfaces. In addition to the difficulty in evaluating

471
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, NOVEMBER 1972

the contribution of these films to the phase angle of the ca- pacitor, the stability of such films must be determined by

periodic reevaluations.

It is with regard to thin dielectric films that cross capacitors were found to have unusual phase angle characteristics. Specifically, it was found that if the addition of a thin dielectric film to one of the electrodes produced an increase in the phase

angle of one of the cross capacitances, it correspondingly pro- duced a decrease in the phase angle of the other cross ca- pacitance. With certain types of cross capacitors, and under

proper conditions, it was found that thin dielectric films on

the electrode surfaces tend only to produce equal and opposite

contributions to the phase angles of the two cross capacitances. A cylindrical cross capacitor that comes close to meeting these conditions is illustrated in Fig. 1. One limitation is that films near the defining ends of the capacitor may not behave as described unless electrical measurements are made

before and after a mechanical manipulation of the electrodes. The toroidal cross capacitor shown in Fig. 2, having cross capacitances CjB and Coi, is not subject to this limitation

since it has no defining ends. Although the cancellation of

film effects with this capacitor is not as complete as with the capacitor shown in Fig. 1, the cancellation is of a sufficient degree for this type of capacitor to be suitable as a standard of

phase angle for precision air capacitors.

Two capacitors of the type shown in Fig. 2 have been constructed and are presently under investigation. Results thus

far indicate that the contribution of thin films to the phase

angle of the average cross capacitance is probably between

one and two orders of magnitude below the bridge sensitivity of ±1 X 10"8 rad. All work has thus far been limited to 1592 Hz.

II. Cylindrical Cross Capacitor

Let the two cross capacitances in Fig. 1 be designated as CAc and CBD , their sum as CT, and let CAc

= Cbd = Q> when

physical symmetry is present and dielectric films are absent.

It has been well established [1] , [2] that if one of the elec- trodes is displaced slightly so that CAc # CBD , then

-ELECTROSTATIC

SHIELD

Cj - 2Cq j + ln2 ( CAC -CBDy +

8 \ CQ

(1)

It has also been established [3] that if a thin dielectric film

is added to one of the electrodes so that CAc ¥= CBD , then (Cac ~ Q?) * ~{CBd

~ Co)- By speculating that the secondorder term contained in(l) might also be applicable to changes

produced by thin dielectric films and by including the loss component of the dielectric, (1) can be rewritten as

rr = 2(/coC0)

1 + ln2 ( YAC - YB0

jcoCo

which yields

In 2 / CAc

- CBd

(0ac ~ 6bd)>

(2)

(3)

Fig. 1. Cross section of cylindrical cross capacitor constructed from

four bars.

Fig. 2. Cross section of a toroidal cross capacitor.

where 8 T, 6AC, and 6 BD are phase angles corresponding to CV. CAC , and CBD . The validity of (3) was checked by applying various types and thicknesses of dielectric materials to the electrodes of a capacitor of the type shown in Fig. 1. A sampling of results obtained for one thickness of vinyl tape is given in Table I. Agreement between measured and calculated values of 6 T is within the uncertainties of the measurements. The following points regarding Table I should be noted:

1) as predicted by (3), 6 T is small when either (CAc

- Cbd)ICo or ®ac ~ ^ bd is small; 2) discontinuities at the ends

of the tape required relatively small corrections; 3) {CAc

- CBD)/Co is independent of its source, i.e., the tape on elec- trodes A and B contribute approximately 0.04 and -0.04,

respectively, to (CAc

- CBd)ICo', deviations from these values

result from deliberate changes in the physical symmetry of

the electrodes; 4) (3) is applicable to relatively thick dielectrics, i.e., t = 0.38 mm = 0.012 r where f is the tape thickness and r is the minimum distance between electrodes A and C;

5) with greater thicknesses of vinyl tape, a third-order term in

t can be observed as differences between the measured and

calculated values of d T; 6) with materials having different

dielectric constants, the third-order term becomes significant

at somewhat smaller or larger thicknesses. Dielectric constants of the materials measured were between 1.1 and 5.8; thick- nesses were between 0.1 and 2.9 mm.

It is also possible to calculate 6 T directly from the dielectric properties. The equations needed for these calculations are derived from basic equations for CAc and CB£, when dielectric films are absent, i.e., CAC = C0 U+k1 t + k2 t

2 + ] (4)

472
SHIELDS: PHASE-ANGLE CHARACTERISTICS OF CROSS CAPACITORS

TABLE I Comparison of the Measured Value of 0 t With the Value Calculated from (3), 8 T = Qn 2/4) HcA c- cBD )ic0 ] (eAC -eBD )

In 2 (£)«">

With

Vinyl CAC ~ CBD 9AC <>BD Measured Calculated Tape Co (prad) (Mrad) Orad) (Miad) A +0.0402 +466 -472 +6.5 +6.5 B -0.0384 -448 +442 +5.6 +5.9 B +0.0638 -425 +434 -9.3 -9.5 B -0.0048 -442 +441 +0.6 +0.7 AaxiAB +0.0304 + 2 -1 +0.2 0.0

Cbd =C0 [l ~ kxt- (*a " k\ In 2)t2 + ]. (5)

where t is the thickness of a metal film on electrode A. If the metal film is replaced by a dielectric film having a thickness t and a dielectric constant e, then CAC = C0 {l+k1 t[fl (e)] +k2

t

2 [Me)]+---} (6) CBD =C0 {1- fc,r[/2 (e)] - (k2 - k\ In 2) t

2 [/4 (e)] +•••}.

(7)

It has previously been shown [3] that from theoretical considerations fi (e) = /2 (e) = [1 - (1/e)] provided that t is small compared with the length of the lines of force terminating at the dielectric. Experimental results indicate that the preceding can be extended to the second-order terms so that /3(e) =

/4(e) = [Me)} 2 = [f2 (e)} 2 = [1 - (1/e)] 2

. If t[l - (1/e)] =

i - jt", then GAC

'AC ~ 1 ~~~ ^ CQ [l+ ki (/ - jt") + k2

(t' - jt")2

] CO (8)

/ ^2 « C0 [l - k x {t' - jt") - (k2 - k\ In 2)

(t'-jt")2 ], (9)

where Gaq an<l GBD are the effective parallel conductances. Thus, to second order, a thin dielectric film on the preceding

capacitor can be replaced by a metal film of thickness /' and

an imaginary component of equivalent thickness t"

.

Separating real and imaginary parts yields

(10)

Gac ^ 3CAc „ bt' GBD _ bCBD „

bt' (11)

Within limitations, a general form of (10) and (11) has been

found to be applicable to other capacitors, i.e., G

CO

bC

Yt'

(12)

dimension. Thus,

6 T (In 2) k\t't" (13)

and when physical asymmetry is accounted for

In 2 -J-Wit' + 2k3 T)(2k 1 1"), (14)

where t is the displacement of electrode A from its position of symmetry. A comparison of (14) with (3) shows that (CAC - CBd)IC0

= (2k, t' + 2kz 7) and (6AC - d BD) = {2k, t"). The preceding relationships were checked experimentally

using a variety of dielectric materials. Agreement was within

the controllability of the various materials. Values of t' and

t" were determined experimentally using a guard-ring parallel- plate capacitor. In the practical case of clean dry metal surfaces and reasonable asymmetry, t' can be neglected since t > t' . Assuming that

t is adjusted to ±5 X 10"6 m, (14) yields

d T/t" * ±1 X 10" u rad/equivalent nm. (15)

where t relates to an actual or effective change in electrode

Values of t" for a variety of metal surfaces were determined by

Astin [4] , i.e., a microradian-millimeter as listed by Astin is equal to 0.5 equivalent nanometers on each surface. Most dry

metal surfaces are listed below 1 /irad-mm. Thus, the magnitude of 6 T can be expected to be less than 1 X 10~ u rad. Since the other three electrodes are likely to have similar

films, the resultant magnitude of Q T can be expected to be

further reduced. As mentioned earlier, the preceding results are not directly applicable to films near the defining ends of

the capacitor.

III. Toroidal Cross Capacitor

Two capacitors of the type shown in Fig. 2 have been constructed of stainless steel. Each of the main electrodes, /, O, T, and B, is supported by two glass balls and one glass spacer. The assembly is suspended from the top plate by three straps and enclosed by a vacuum housing.

The preceding contains three important capacitances: Ctb,

Cqi, and CTj. CTB and Coi are 0.5-pF cross capacitances.

Cji is a 10-pF capacitance, which serves essentially as a means

for evaluating t". Let Ctb + Coi = Cs 3110 iet ^rs> ^o/> ^s. and 8 Tj be the phase angles corresponding to CTB , Coi, Cs , and CTj. The values of d TB/t", 8 OI/t", and 6s /t" listed in Table II are applicable when dielectric films are thin and (CTB ~ Cqi)I

Cs ^ 1. Values were determined theoretically using (12) and

checked experimentally with dielectric materials as previously

described. The contribution of second-order terms to 6 S was found to be approximately the same as with the preceding cylindrical cross capacitor and can be neglected when clean metal surfaces

are present. Thus, from Table II

9 s lt" « 3 X 10"9 rad/equivalent nm (16)

assuming that identical films are present on all electrodes. In a

473
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-21, NO. 4, NOVEMBER 1972

TABLE II Effects of a Standard Film on the Toroidal

Cross Capacitor's Electrodes

8tbI'" S OIlt" 6Slt" (n rad/ (n rad/ (n rad/ Electrode equivalent nm) equivalent nm) equivalent nm)

T + 113 -112 +1 0 -127 +104 -12

I -103 +130 +13 B +113 -112 +1 Total -4 +10 +3

similar manner it was found that

6 TI/t" « 6 X 10" 7 rad/equivalent nm. (17)

The significant fact here is that 8 TI/6 S « 200. Combining this ratio with the value of 6 TI - 6S

obtained from a bridge comparison of CTi and C$ yields values for the true phase angle of each capacitor.

It is assumed in the preceding calculations that uniform films

are present on all active electrode surfaces. Since it is unlikely

that 6 77 will be more than one order of magnitude above the estimated uncertainty (±2 X 10"8

) in the measurement of 077 ~ 0£> a substantial allowance can be made for possible

nonuniform films with no significant reduction in accuracy.

IV. Conclusion

Thin dielectric films on the line or detector electrodes of a three terminal capacitor tend to produce positive contributions to the phase angle; thin dielectric films on the grounded

electrodes tend to produce negative contributions. The sym- metry associated with cross capacitors indicates that thin

dielectric films should produce approximately equal and opposite contributions to the phase angles of the two cross capacitances. This has been confirmed experimentally as pre- viously described.

It is planned that the two toroidal cross capacitors will be- come permanent standards of phase angle. Sources of error as- sociated with the realization of these standards will be de- scribed in a future paper.

References

[1] D. G. Lampard, "A new theorem in electrostatics with applications

to calculable standards of capacitance," Proc. Inst. Elec. Eng. (London), mono. 216M, Jan. 1957.

[2] A. M. Thompson, "The cylindrical cross capacitor as a calculable standard," Proc. Inst. Elec. Eng. (London), vol. 106, pt. B, May

1959.

[3] D. G. Lampard and R. D. Cutkosky, "Some results on the cross capacitances per unit length of cylindrical 3-terminal capacitors with thin dielectric films on their electrodes," Proc. Inst. Elec. Eng. (London), mono. 35 1M, Jan. 1960

[4] A. V. Astin, "Nature of energy losses in air capacitors at low fre- quencies,"/ Res. Nat. Bur. Stand., vol. 22, June 1939.

474
I

JOURNAL OF RESEARCH of the National Bureau of Standards — C. Engineering and Instrumentation

Vol. 74C, Nos. 3 and 4, July-December 1970 Techniques for Comparing Four-Terminal-Pair

Admittance Standards

R. D. Cutkosky

Institute for Basic Standards, National Bureau of Standards, Washington, D.C. 20234

(June 26, 1970)

Some of the advantages of four-pair admittance standards and some of the special problems

encountered in their measurement are pointed out. Detailed descriptions of three distinct types of four- pair bridges and some of their limitations are presented. These three bridges form a vital part of a very

precise absolute measurement of resistance based on a calculable capacitor being undertaken at the National Bureau of Standards, but are believed to be of more general usefulness.

Key words: AC direct-reading ratio set; ac measurements: bridge: coaxial chokes: defining trans- formers: equal power bridge: four-pair standards: frequency-dependent bridge: quadrature bridge.

1. Introduction

The quest for improved accuracy in the measure- ment of audiofrequency admittances has led from the now universally accepted practice of utilizing two pair

standards, such as so-called three-terminal capacitors

for high-impedance devices and such as mutual inductors for low-impedance devices, to a composite system

combining the advantages of both techniques. A theoretical justification of this composite system in which

the standards are provided with four pairs of terminals was published in 1964 [1] ', but with the exception of a paper by Homan [2] which was limited to measure- ments on fairly small impedances, no systematic in- vestigation of four-pair admittance measurements has

yet been described. The process of converting to four- pair measurement systems has been underway at the

National Bureau of Standards for several years, during which time many of the special problems involved

in their use have been investigated. This paper is an

attempt to describe some of the techniques which have

evolved, and to indicate the order of precision that may

be obtained.

The techniques described in this paper are of general applicability but were developed for the specific

purpose of comparing the reactances of capacitors

having nominal values of 10 pF with the resistance of a

special 1000-fl resistor having a negligible frequency

dependence [3J. The three basic types of bridges used

for making this comparison are described here in some detail. It is intended to calibrate the 10-pF ca- pacitors by means of a calculable capacitor now under

construction at NBS. A dc comparison of the 1000-fl

resistor with the bank of 1-Q resistors presently maintaining the NBS unit of resistance will then complete

an absolute measurement of the ohm.

Because of the large number of intervening steps

between the calculable capacitor and the bank of

1-H resistors, very elaborate precautions are necessary

to maintain a high level of accuracy. Using the tech- niques outlined here, it is expected that the principal

uncertainties in the measurement sequence will be

due to mechanical imperfections in the calculable

capacitor.

Although a rather complex sequence of auxiliary

balances is required with the bridges described in this paper, they are not prohibitively time-consuming for the limited number of comparisons needed in an

absolute ohm determination. These techniques would

be considerably less attractive for a bridge in constant

use, but this would be less of a drawback in a computeroperated system. It is considered that techniques

similar to those described in this paper are essential

if uncertainties must be kept smaller than 1 part in 108 over a large range of impedances at audiofrequencies.

The equivalent circuit of a four-pair admittance

standard is shown in figure la, taken from the paper

cited above [1], and in an abbreviated form, in figure

lb. The standard may be viewed as a device in which

the ratio of current in pair 3 to the open circuit voltage

at pair 2, subject to the condition that both the voltage

and current at pair 4 are zero, is the parameter of

interest. This parameter has the dimensions of admit- tance and may be written

pu = ±l i> = i. i = 0, e4 = 0.

1 Figures in brackets indicate the literature references at the end of this paper.

It may be shown using figure la that pk=y,(i+K2z,)(i+nz3 ).

(i)

(2)

475
An interesting reciprocity theorem also exists, which

shows that P-A = P

4A f° r anv four-pair network.

3)3

FlGL'RE 1. Four-pair admittance standards: (a) Complete circuit.

lb) Pictorial representation.

Some properties of four-pair admittance standards

which make them particularly useful are their relative

insensitivities to variations in series impedances and

shunt admittances in the leads as is implicit in (2), and the fact that standards of either very small or very large admittance may be constructed in such a way that their four-pair admittances are nearly

identical with their ordinary two-pair admittances

obtained by either open circuiting or short circuiting

the two extra terminal pairs. These properties are discussed in the reference cited above [1]. Ordinarily the relationship between the four-pair

admittance of a standard and its admittance when

treated as a two-pair standard is of interest only when

the standard is some kind of absolute or calculable

instrument, and it is necessary to determine the

difference between the calculated admittance at some

internal and inaccessible location, and the effective

four-pair admittance at the external connection points.

Examples of this type of standard are calculable

capacitors, calculable inductors, and resistors having

calculable phase angles or ac-dc differences. Such

devices are outside the scope of this paper, which is restricted to the problem of comparing four-pair admit- tance standards with each other. Implicit in the adoption of either a four-pair or a two-pair approach is the requirement that the connections are indeed treated as pairs, and that there is no net current from one terminal pair to another. Many

advantages result from grouping the terminals of a standard in pairs. Consider the simple case of a three- terminal capacitor provided with two coaxial connectors

and with a permanent connection between the shields

of these connectors. Conceptually the measurement

of the capacitor consists of applying a voltage, e, to one connector pair and measuring the shoit circuit current i at the other connector pair. The transfer

admittance of the device is then given by

Y = ja)C + G = L.

The transfer admittance defined in this way is a function of the impedance in the connection between the

shields of the coaxial connectors, but there is no ambiguity in the definition as there would be if a single

ground connection of unspecified placement were used

for both the input and output.

One of the principal reasons for defining the standards

in terms of isolated coaxial connector pairs as sketched

above is that under these conditions there is exactly

the same current but with reversed sign in the inner

wire of each cable as there is in the surrounding

coaxial shield, and therefore the magnetic field exterior to both cables is zero. This eliminates possible

mutual inductive couplings between the cables used

to interconnect a number of components together to form a bridge. Likewise, even if stray magnetic fields were present in the vicinity of the coaxial cables, no

induced voltages would be produced by these fields

to affect the measurement. Although these arguments

pertain only to the connecting leads of the devices,

the devices themselves can be quite easily isolated

from each other by providing them with individual

mu-metal shields when this is needed.

Four-pair admittance standards are constructed

with four coaxial connectors, and the advantages

gained by preventing net currents from flowing

between the four connector pairs are very great, as

this eliminates external magnetic fields and their

effects on the measurement. In this respect, the

practice of dealing with five terminal standards having

a single, common ground connection cannot be

recommended.

It is necessary to maintain the conditions of zero net currents in the cables of components defined as outlined above when they are interconnected to form a bridge circuit. Coaxial chokes were developed for dealing with this problem [4], and are effective in attenuating the net currents in the cables by a factor

of several hundred. Unfortunately the net currents are

not completely suppressed by coaxial chokes. A method

for estimating the errors caused by incomplete sup- pression of net currents by coaxial chokes is described

later in this paper. Most of the bridge circuits appearing in this paper

show explicitly the coaxial nature of the components.

This was done to emphasize the importance of considering the return currents in the shields. After some

practice in dealing with circuits drawn in this way,

they are much easier to relate to the physical bridge

setup than are the conventional textbook bridge cir- cuits. Some difficulty may be experienced at first in translating the still useful and important body of

classical bridge theory to the coaxial form of presentation. It may be an aid in understanding some of the

circuits in this paper to redraw them with a common

ground point. If this is done, the similarities between

many of the auxiliary balance systems used here and

common classical techniques, such as yoke and lead

balancing in Kelvin double bridges, and the use of

Wagner and conjugate Wagner arms in ac bridges, can be readily seen. Redrawing the circuits in this way must be thought of purely as an aid to understand476
ing, because any attempt actually to construct a cir- cuit without proper attention to the details of the shield return circuits would result in large and erratic errors dependent upon the accidental arrangement of the

leads. A final but important introductory point about comparisons of four-pair admittances, or indeed any other

kind of admittances, is that even though the standards

are defined subject to certain constraints on the cur- rents and voltages, it is not necessary when comparing

two such standards to satisfy any of the defining conditions: it is merely required that the same result be

obtained for the ratio of the four-pair admittances being

compared as would have been obtained if the conditions had all been satisfied. This observation opens up

a very wide range of possibilities in the design of bridge

circuits, and is related to the techniques described by- Thompson for dealing with bridges involving multiple

balances [4]. The Kelvin bridge for comparing four- terminal dc resistors also makes use of this principle,

since currents exist in the potential leads of the resis- tors, but cause no errors.

2. Basic Comparison Circuits

Figure 2 shows four four-pair admittances connected

in a bridge circuit involving six generators and six detectors. The balance procedure consists (for ex- ample) of arbitrarily fixing one generator, and then

adjusting the other five generators and one of the four

Fi<»L RE 2. Elementary four-pair bridge.

four-pair admittances until all six detectors are bal- anced. The balance condition is P\P^ = P: P<,. If all generators and detectors are interchanged, the bridge

network retains its form, and it may be shown, using

the reciprocity theorem for four-pair networks, that the balance condition is unchanged.

The scheme of figure 2 contains a number of drawbacks, chief among which is the difficulty in simultaneously nulling six detectors,- all of which are affected

in various ways by the six adjustable parameters. This

problem is solved in principle by mixing the various

null detector responses in a suitable combining net- work to obtain six new null detector responses, each of

which is a weighted average of the six original null detector inputs. The combining network parameters

are chosen so that the matrix relating the new detector

responses to the adjustable parameters is diagonal

[4.5]. In this case, only the detector which responds to the adjustable four-pair admittance, called the main

detector, is of interest, and the others need to be bal- anced only if the non-diagonal elements of the matrix

are not exactly zero. A more fundamental problem encountered in realiz- ing figure 2 has to do with shunt admittances to ground

associated with the detectors which compare the open

circuit voltage of one standard with that of another, and

with shunt admittances to ground associated with the

generators between the current leads of two adjacent

standards. By the reciprocity theorem the effects of

these sets of leads are similar, and any solution found

for one problem may be immediately applied to the

other. The nature of the problem and the order of magnitude of the possible errors likely to occur may be seen from (2). which shows that the effect of admittance to ground in the potential lead is to load the leads joining

the internal junction points of the standard with the

point at which the open circuit potential is to be measured. The effect of admittance to ground at the cur- rent lead is to shunt some of the current to ground

before it enters the adjacent standard.

The lead effects described above are unlikely to ex- ceed a few parts in 10" at audiofrequencies, unless

interconnecting cables longer than several meters are

used. Even with very long cables, no errors result pro- vided that these cables are treated as part of the standard, so that their effects are the same when the

standard is calibrated as when it is used for calibrating

another standard. However, the stabilities of the series

impedances and shunt admittances of the cables must

be considered.

The network used to interconnect the ends of the

two cables and the null detector or generator is in principle a three-pair network, since the interaction of the

null detector or generator terminal pair must be in- cluded in the matrix description of the network. This

network must be- carefully constructed so that the

admittances shunting the cables from the standards

are small, equal, and measurable: and so that the inter- action of the null detector or generator with the other

two terminal pairs is simple to analyze. An ideal threepair circuit for accomplishing the desired objectives

would provide a direct, completely reproducible con- nection between the two coaxial cables, except for an

infinitesimal gap between their inner conductors, and

with provision for connecting a null detector or generator across this gap. A network electrically equivalent to the above is employed at NBS which we call a defining

transformer, shown pictonaily in figure 3. A particular

advantage of this network is that one side of the detector or generator terminal pair is at ground potential.

When used in the potential leads of the bridge

shown in figure 2, the potential leads from the two

standards are connected to terminals A and B of the

defining transformer. The direct internal connection

between the inner conductors, in connection with the

477
return through the outer shield, provides a one-turn

loop around the high-permeability core shown shaded.

A detector is connected to a 100-turn winding on this core via the coaxial connector C, and indicates a null

FIGURE 3. "Defining transformer" for terminating current and

potential leads offour-pair standards.

if the potentials at A and B are equal. Capacitance

between the 100-turn winding and the connection

between the inner connectors of.-/ and B could cause

substantial errors. An internal shield with a thin air gap as shown eliminates this problem. Incomplete

shielding can be detected by interchanging A and B. When using defining transformers in a bridge, the

potential leads are considered to terminate at the center of the device, in the region of the insulated

gap in the inner shield. A number of nearly identical

defining transformers have been built such that they

may be used interchangeably to terminate the potential

leads of the standards without changing the loads on

the leads. These devices are used in the current leads

also, in which case a generator may be substituted for the null detector. Some objection may be made to the use of these

devices, in that the standard is no longer an entity in

itself, but has a definition dependent upon a termination; which might more properly be considered as part

of the measuring instrument. Precedents do exist for

this type of procedure, an example being the use of a precision coaxial connector on a two-terminal capacitor,

for which a correct measurement requires a mating

precision connector. In practice a very substantial

simplification of our measurements was found to result from the introduction of defining transformers.

The technique which we use for measuring the

effects of net currents between one terminal pair and

another in a bridge containing coaxial chokes is rather

cumbersome, but provides a reliable measure of the

errors caused by non-ideal coaxial chokes. It is first assumed that every ground loop in the system to be

investigated contains a coaxial choke, and that the

minimum possible number of chokes to accomplish

this purpose is used. This does not imply that one

cannot add as many ground loops as may be desired,

but simply that in the final circuit all ground loops are interrupted with a minimum number of coaxial

chokes. Singling out a particidar choke for investigation, a single turn of wire is wrapped around the core

of the choke and connected to an auxiliary detector.

The deflection of this detector is a measure of the

voltage tending to drive current through the choke.

A quantitative measure of this voltage may be

obtained using the voltage insertion transformer shown

in figure 4. This device consists of a high permeability

FlCLRE 4. Voltage insertion transformer for investigating errors caused by net currents in coaxial cables.

toroidal core wrapped with 200 turns of wire connected

to a coaxial connector. A toroidal mu-metal shield

surrounds the core and winding, so that negligible

magnetic fields exist outside of the shield when the

200-turn winding is excited. In use, the cable leading to the choke selected for testing is passed through the

hole in the shield of this device, forming a 200:1

transformer. A voltage source derived from the same

oscillator supplying the bridge, and adjustable in both

real and imaginary components, is connected to the

200-turn winding. We find that an operational amplifier

circuit described in an earlier paper [6] is convenient

for this purpose.

The voltage source is adjusted until the auxiliary

detector monitoring the voltage across the coaxial

choke registers a null. The real and imaginary parts

of the adjustable voltage required to obtain this null are recorded. The next step is to remove the auxiliary

detector and to change the adjustable voltage by an amount large enough to yield sufficient sensitivity

but not large enough to drive the choke into a nonlinear region. The real and imaginary parts of the

change in balance point of the bridge caused by this change in applied voltage are recorded. A straightforward manipulation of the various complex quantities

obtained by this scheme allows one to calculate the

real and imaginary parts of the error caused by in- complete suppression of net current by the coaxial

choke.

All of the coaxial chokes in the system are evaluated

in this way. An algebraic addition of the individual

errors then gives the total error, which is applied as a correction to the raw results of the measurement.

The technique sketched above could be generalized

for a system with M chokes by providing each of the

M chokes with voltage insertion transformers, and by

bringing all of the choke voltages simultaneously to a null as indicated by M auxiliary detectors. In this case no net currents with their attendant errors would

exist, and there would be no corrections to apply to the bridge balance. The complete system applied in the circuit of figure 2 could then be viewed as one with

478
4/ -1-6 adjustable parameters and with V/ + 6 null detectors. One could conceivably construct a combining network to diagonalize the matrix relating these

two vector quantities to make the effects of net choke

currents less important. This has not yet been required

for our purposes, since the residual errors due to net currents through the chokes are small enough to be

measured with adequate precision using the extrapolation technique sketched above.

It is very important that one uses the minimum

number of chokes necessary to break all ground loops

when the normal loop voltages are measured by balancing them out as indicated above. If two chokes of equal

impedance were in series in a particular loop, the

method described would assign the total loop voltage

to each choke, when in reality each choke would have

across it only half of the total voltage. A simple test for multiply choked loops is to measure the voltages

at each choke produced by exciting each loop in turn with a voltage insertion transformer. If the nondiagonal terms of the resultant matrix are much smaller

than the diagonal terms, no multiply choked loops exist.

It is found in measuring the effects of the chokes in a given system that they can often be placed in cate- gories according to the amount of voltage driving the

loop, and according to the effect of a loop voltage on

the bridge balance. The measurement error caused by

a coaxial choke having insufficient impedance is large

only if the loop voltage driving a choke is large and

if a voltage in this loop has a strong effect on the

bridge balance. It is usually possible to arrange a bridge in such a way that no choke has both un- desirable characteristics. This usually requires the

addition of extra ground connections with an extra choke for each added connection. The procedure for accomplishing this is at this time largely empirical.

Coaxial chokes do not provide the only means

for suppressing net currents in coaxial cables. Isolation

transformers are used in some cases, and net currents can then result from unbalanced interwinding capacitances. If the interwinding capacitances are very

poorly balanced, the effective ground loop voltage can be quite high, and the accurate measurement of the

errors due to the resultant ground loop current is rather difficult. In such cases it is usually best to measure the current directly, for example by converting

the voltage insertion transformer to a current sensor by connecting a phase-sensitive voltmeter in parallel

with a low impedance load to the 200-turn winding.

About the only other errors likely to occur in a fourpair bridge are those caused by stray couplings between

the various components. These effects can usually

be made negligible with electrostatic and electro- magnetic shields, and tests for the effectiveness of

these shields are not difficult to perform. Eccentric

coaxial cables usually cause no serious problems, but

must be considered. In some cases acoustic couplings can be troublesome, particularly if the generator and

detector circuits both contain partially magnetized

transformers. The 100:1 bridge described in section 5 suffered from this effect initially, but after the cores were carefully demagnetized and after the addition of

some acoustic insulation, the errors due to acoustic

coupling were reduced to a few parts in 10 10. The

mechanism responsible for the acoustic coupling

involves magnetostriction in the core of the transformer

in the generator circuit. If the core is partially magnetized, the acoustic output of the core will contain a component at the same frequency as the generator in addition to the expected but less troublesome components at the even harmonics of the generator

frequency. The resultant mechanical excitation of the

transformer in the detector circuit causes no trouble

unless the detector transformer is also partially magnetized, in which case a fundamental frequency component of current can be generated.

Some care must be taken when adjusting the meters

of the null detector to zero deflection in the absence

of a signal. It had been the practice at NBS to discon- nect the admittance standards of a bridge from the

voltage transformer to which they were normally

connected and to zero the meters with the voltage

transformer excited. This technique eliminates most

of the errors due to magnetic coupling from the

oscillator and ^voltage transformer to the detector,

and also eliminates some of the errors caused by imperfect coaxial chokes. When the choke errors are measured independently as sketched above, this procedure could result in a double correction for some of the choke errors.

3. Quadrature Bridge

As the first example of an actual four-pair bridge, we will use a type of frequency-dependent bridge of which two-pair versions have been in existence for some time [4. 7. 8. 9]. The circuit is basically the same

as that described by Thompson [4] and uses the twin- tee detector coupling network proposed by him. The

NBS bridge contains two 105 -Q four-pair resistors and

two 1-nF four-pair capacitors, and operates at an

angular frequency of 104 rad/s, which corresponds to a frequency of about 1592 Hz. Figure 5 shows a partial

schematic of the NBS four-pair quadrature bridge, in which D| through Z) s are null detectors. T:-, through T» are defining transformers, and P\ through P4 are the

four-pair admittance standards. Seven of the eight

complex adjustments required to balance the eight

null detectors are indicated by either admittances or generators with arrows through their symbols. These

seven adjustments are required only for realizing the

defining conditions for the four-pair admittances, and

may be uncalibrated. The eighth complex adjustment

required to balance the bridge is made through the use of two seven-dial inductive voltage dividers T-> and

T,u which drive small fixed capacitors labeled and

y«. This complex pair of adjustments serves to indicate

the relationship between P\. P->, P:t , and P*. Using the

notation indicated on figure 5, we have when all detectors are balanced

i\ = etlPt + eaYr, = — e c P> and

i>^ebP,+ efY^-ecP:i .

479
lOOkfl

FlCLRE 5. Four-pair quadrature bridge with detector combining network deleted.

If we represent the ratio of T4 by e& = — e„(l + 8)

with 5 complex and | 8 | <§ 1 . and since ea ea (2Ai— 1

)

and e/ = — e& ( 2A>— 1 ) . we have the balance condition

P3 [/

>

i + n(2A 1 -l)] =-P2 (l + 8)[P-y6 (2A-2-l)].

(3)

A repeat measurement is made with the connections

interchanged at points A and B. and with the connections between T4 and the defining transformers T5 and

Tq interchanged. This reverses the sign of 8 to first order, and also reverses T> and 7^, so that

e'd = ea (l — 2A,') and = — e&(l — 2^,).

The resultant balance condition is P3 [Pi + y-Al-2k[)] --P2 (l-8)[P4 -r6 (l-2A;)].

(4)

Averaging (3) and (4) we have

PAPi + Yiih-ki)] ~-Pt [P<-Ye{k2-kl)l (5)

If we let Pn = Gn+jcoC„, Y-0 =jwC-0 and Y6 —j(oC6 and assume that Gi = G3 = (dC> = a)C4 and that a»Ci <t d,

a»C:i <^ G3, Gi^toC' and £4 <§ o>C4 . we may separate

real and imaginary parts of (5) to obtain

G%G\

2 C2 C4 = 1 c4 (fa-A-i) (6)

which relates the main components of the four-pair

admittances, and

GI ~1~ /-» 1 ' ^ (ffl # 1 ^

1 Lr.i (1)L> 0)La

(7)

which relates the phase angles of the four-pair

admittances.

The approximations made in deriving (6) and (7) cause no errors exceeding 2 parts in 1010 for the NBS

four-pair quadrature bridge. Under less ideal conditions, the second-order correction terms might be

required.

480
The nominally 1 : 1 transformer in figure 5 is of

two-stage construction [6, 10], with a mu-metal shield

between the first and second stages, and with another

mu-metal shield surrounding the entire transformer. It can be seen that when the null detectors D& andDj are balanced, the output terminal pairs of T4 are subject to open circuit conditions as defined at the central refer- ence planes of the defining transformers T-a and T$. The

voltages e<2 and e& in the equations above are referred to these reference planes in a manner analogous to the

definitions of the open circuit voltage.3 of the four

pair standards. Reversal of T4 is accomplished by re- connecting the cables at the points of entry to T5 and

T9, which has a negligible effect on the open circuit

ratio. Three of the eight null detectors shown in figure 5

are not operated on by the combining network used

with the bridge. The detectors labeled Z) 8 , Z) 7 , and D*

connected to T^, T7- and are brought to a null one

at a time by adjusting Y7, Y\o, and either Y* or Yj, after which these detector terminals are shorted to reduce

the effect on the main detector of a slight error in these

auxiliary balances. This technique works very well for adjusting Y7 and Yio, but some convergence problems

exist with the balance of Yh or Fa. Ideally, one would

like to null the detector Z) 8 connected to T-, by adjusting

Y»— Y» whiie keeping F« + Y» fixed. One of the other

balance conditions turns out to involve Y$+ Y» but

not Y*—Y9 . In practice, we have been able to achieve

fast convergences of these two auxiliary balances

without the elaboration of ganged switches by manually

tracking Yg with Fy. The guiding principle behind the combining net- work used with the bridge (see fig. 6) is to combine

the various detector terminal pairs two at a time and

further to combine the new detector terminals so formed until only one detector terminal pair remains.

We begin by combining the detector terminals Di and

D> of Pi and P> to form a single detector terminal D»

which does not respond to excitation of T6 , and by

similarly combining the detector terminals and

of Pi and P4 , to form a single detector terminal Di0 which does not respond to excitation of T$. The initial combining network linking the detector terminals D\

and D> can be understood by letting ea = ec — 0 in figure 5 and observing that the circuit between Ts and Z) H in figure 6 is a Schering bridge, for which a complex

balance may be obtained by adjusting the two variable

capacitors C n and Cv>. The circuit is adjusted by

actually connecting the bridge generator to the input

lead of TH rather than to the input of T\. After C n

and C\-> are adjusted, the input terminal of 7^ is shorted.

The result is that although the voltage at neither Di nor D> is exactly nulled, these voltages are very small; and since changing the excitation on Ta does not change

the voltage at D9 , no error results from failure to adjust

this excitation to obtain individual nulls on the detector

pairs D\ and D>. The combining network between the

detector terminals of and Z> 4 is identical with that described above, and is adjusted by varying C, 3 and

C14. The relative sizes of the admittances are chosen

to produce a minimal effect on the bridge sensitivity

and on the operation of the twin-tee combining network

Unking Z)9 and Di 0.

D4 FIGURE 6. Detector combining network for quadrature bridge.

The network linking Dg and D10 may be understood

by letting all generators in figure 5 be zero and by

assuming that a generator is inserted to make ec ^ 0. The network can be recognized as a twin-tee circuit

in which a null can be produced at Du by adjusting

Rxs and C>i. After making this balance, the voltage at Dn is independent of the auxiliary adjustment Yg + Yg. In practice the twin-tee is adjusted so that changing

Yg + Yg produced no change in the detector voltage

at Du and Yg + Y9 is adjusted so that changing the

twin-tee setting produces no change in the detector

voltage at Du - Although a twin-tee combining network is convenient

for a quadrature bridge, a detailed noise calculation

shows that it rather greatly augments the thermal

agitation noise already present in Pi and P:i . With 4

mW dissipated in Pi and P3 , one has difficulty in detecting an unbalance smaller than one or two parts

in 109

. A substantial improvement could be made by

maintaining the combining network at low temperature,

or by separately amplifying the voltages at D9 and

Dio before coupling them together.

The final combining network links Du and D5 with

Dn. It is adjusted by means of the decade inductive

voltage divider Ao and the associated phase-shifting

network so that inserting a 5-il resistor in one of the

cables joining 7\ with T-z, T3 , and T4 to simulate the

generator shown in figure 5 does not change the

detector voltage at D)2 . After making this adjustment

the 5-H resistor is removed.

481
The sequence of adjustments outlined above is easier to perform than to describe. The entire bridge can be adjusted in 10 minutes, and the auxiliary ad- justments have been found to drift in a week by less than that required to yield an error of 1 part in 109

. The complete quadrature bridge consisting of figures 5 and 6 contains 25 coaxial chokes, each of which was checked using the techniques described in section

2. The largest individual error was found to be 1 part

in 1010

. and the total error resulting from the existence

of net currents in all chokes, the algebraic sum of the

individual errors, was about 2 parts in 1011. Some special problems are encountered in a quadrature bridge simply because the balance is frequency-dependent. One especially important

problem is caused by intermodulation distortion in the bridge detector between adjacent harmonics of the

bridge fundamental frequency. These distortion

products may have components at the fundamental

frequency, which would cause a substantial error. The individual harmonic components entering the

detector of a frequency-dependent bridge are not nulled with the fundamental as they are in a frequencyindependent bridge, and may be of rather high amplitude. We use a special filter between Tu of figure 6

and the detector amplifier to reject all harmonics of

the bridge fundamental frequency before the signals

reach any strongly nonlinear elements. The filter contains a bridge network to obtain a zero transfer

admittance for the second and third harmonics of the

fundamental frequency, and a doubly tuned circuit to attenuate all higher harmonics. The circuit is similar

to that used in an earlier quad bridge [7] but has been

modified to yield a smaller noise figure.

The need for a good filter is indicated by the fact that removing the filter results in an apparent change

in the bridge balance of several parts in 106

. With the

filter in. tests indicate that the error due to intermodulation distortion is less than 1 part in 109

. A convenient

check on the effectiveness of the filter is to augment

each of the harmonics present at the detector terminal

Dy> by injecting an additional current into D\- wjth a small capacitor connected to an auxiliary o^iRator

tuned to the appropriate harmonic. Other tests using

two auxiliary oscillators confirm that the effect is indeed due to intermodulation distortion.. The .tech- nique is very sensitive, and -can be used to detect

distortion products resulting from .nonlinear mixing of

frequencies up to at least the iBth-.har^onic, The

technique gives only an upper bound to the error and would probably not be reliable for determining

corrections.

If a frequency-dependent bridge is balanced at an

angular frequency a> = coo, then for o> ~ too* the detector input voltage will be given by eit — k\ {jtff-.&n)e;l where k\ is a complex constant and ey

is the bridge

generator voltage. If the detector contains a..sharply

tuned single section filter (6 dB/octave) having a bandwidth 2a>rf, then for frequencies near coo the detector

output voltage will be of the form

(8)

The power transfer function between the bridge input

terminals and the detector is then given by

TF, * ( CO — CO.i) Aco 2

1 +

CO — COp

COrf

(9)

If a two section filter were used (12 dB/octave). the

power transfer function would be

TF- *

Aco

i + i^V (10)

COrf TFx and TF- are plotted in figures 7 and 8 for several

values of con. In both cases, oscillator noise components

with frequencies very near co0 are strongly attenuated,

and those at the edge of the filter pass band are the

least attenuated.

~ ~~^^

N\(

1 -— v

T 1 1 1 1 1 1 1 1 1 ^ 1 1 1 1 1 1 1 1 1

1

TF,

1.0

0.8

0.6

0.4

0.2

-10-9-8-7-6-5-4-3-2-1 0 I 2 3 4 5 6 7 8 9 10

FIGURE 7. Power transfer functions for a frequency-dependent

bridge followed by a single section filter.

TF2 04 /— CUd

.|

' iT 1 i i 1 1 r i

02

00 -10-9-8-7-6-5-4-3-2-1 0 I 2 3 4 5 6 7 8 9 10

FlGL'RE 8. Power transfer functions for a frequency-dependent

bridge followed by a double section filter.

The total noise power reaching the detector depends

upon the noise spectrum of the oscillator. This is not known a priori, but in general is highest near coo. We

will assume arbitrarily that the noise spectrum is of constant amplitude, in order to simplify the calcula482
tions. In this case, the total noise power with a single

section filter would be of the form

including that due to thermal agitation noise, is believed

to be less than 1 part in 109.

(&> — a»o) 2

1 +

a» — a»n

ion

(ID

and for a two-section filter.

OJ — OJo

1 +

OJ — O>o Jo;

77CD (12:

The expression (11) for JT'i is not realistic because (9)

is correct only for oj ~ coo- In general TF\ would de- crease when \oj is very large and would not remain

constant, so that W\ would be very large but finite. The advantage of using a two-section filter is obvi- ous from figures 7 and 8 and from (11) and (12). and in addition the very strong dependence of W-> on the filter bandwidth can be seen. If the oscillator noise spectrum

had a peak at oj = ojq, the exponent of oj^ in (12) would

be reduced, but in practice the observed detector

noise due to oscillator noise decreases faster than that due to ordinary thermal agitation noise in the bridge

components when the detector bandwidth is decreased.

We find with the NBS quadrature bridge connected to a reasonably clean oscillator that the use of two cas- caded filters with time constants of 0.3 s makes the portion of the detector noise due to oscillator noise about

equal to that due to thermal agitation noise in the

bridge components. Filters with 3-s time constants are normally employed, in which case the effect of oscillator noise is negligible.

The above discussion deals with noise power and

makes no distinction between the different effects of FM and AM noise. If a frequency-dependent bridge is connected to a two-phase phase sensitive detector

adjusted so that the first channel responds only to the

"real" part of the bridge balance and the second

channel responds only to the "phase" part of the bridge

balance, then the effect of FM noise on the oscillator

is to give noise only on the first or real channel of the

phase-sensitive detector, as might be expected. It can

also be shown that the effect of AM noise on the oscil- lator is to give noise only on the second or "phase"

channel of the detector. The latter effect is not usuallyanticipated. One consequence of this effect is that when the power level of the bridge is being changed,

the phase channel of the detector is thrown off balance,

usually by enough to saturate the detector and to require a short wait before the detector recovers. The separation of FM and AM noise with a frequencydependent bridge can be helpful in the study of

oscillator noise. The main components.fi, P>. P3, and P4 of the NBS

quadrature bridge are maintained at 25±0.0U1 °C in an

oil bath. The precision of the bridge is limited by the

temperature variations in the bath and by the load

coefficients of Pi and P3 . The sum of all other errors.

4. Four-Pair Direct-Reading Ratio Set (DRRS)

The bridge to be described next was actually the first four-pair admittance bridge to be completed at NBS.

and is in some ways not as well conceived as the more

recent quadrature bridge described above. It is nevertheless a highly accurate and wide range instrument,

and has received much use in the last few years. The

bridge has a number of rather complex features,

making it difficult to understand without prior ex- perience with four-pair bridges. It is for this reason

that its description follows that of the quadrature

bridge.

The four-pair direct-reading ratio set was intended

to provide a means of comparing two four-pair admit- tances whose ratio is nominally 10:1. It was designed

so that it could also be used to compare ordinary

two-pair standards. Fewer auxiliary balances are re- quired with the two-pair configuration, so this usage

will be described first. An elementary schematic of the bridge is shown in figure 9. The two admittances to be compared are

labeled Y\ and Y>, and are connected to three star connectors as indicated. The star connectors are constructed in accordance with a design first described by

CURRENT LEAP

CORE I

CURRENT ,_EA0

FlGL RE 9. Elementary 10:1 bridge using a two-stage transformer.

Hamon [11] for dc junctions, but are provided with

ground potential shields to allow their use with alter- nating current. The equivalent circuit of an ideal

four-pair star connector consists of five impedances

linking the five terminals to an inaccessable internal

junction point. The impedance to ground is typically

much larger than the other four. With this system, the

admittances under test are considered to terminate at the internal junction points of the star connectors,

rather than at the ends of the coaxial cables leading

to the standards. The discrepancy between the two

483
definitions of a standard is small if its admittance is small, but could be accounted for to first order. The transformer in figure 9 is a two-stage device in which the 25-turn primary and the 55-turn current winding are wound around core 1 only. The 50-turn

and 5-turn potential windings are wound around both

cores and are isolated from the inner windings and the

two cores by a mu-metal shield.

If there were no capacitance between windings or to ground, there would be very little current in the

potential leads to Y\ and Y>, and a detector connected

to the star connector between Yt and Y> would register a null when YJY-> equalled the open circuit ratio of

the potential windings. Because of loading effects,

both in the potential winding circuit and in Y\ and Y< and the associated cables, two auxiliary balances are required to obtain zero current in both potential leads. The first auxiliary balance can be conveniently ob- tained by means of an adjustable admittance to ground

y4 so that Y3 and F4 serve as a conventional Wagner

circuit, and the second balance can be obtained by means of an auxiliary winding around core 2 only (not

shown), whose excitation is adjustable and derived

from a high impedance source. It may be noted here

that one anticipates that only a very small excitation

will be required on core 2, to drive current through the

interwinding capacitance of the transformer potential

windings, and that very little voltage will appear across

the auxiliary winding on core 2 because of the two-stage

construction. Ideally, an adjustable current source

could be used to drive core 2. The balance procedure for the system of figure 9

would be first to adjust Y\ or Yz to achieve a null on the

detector: second, to disconnect both potential leads

between the transformer and the star connectors and

to adjust Ki to restore the null (a balance which is completely independent of the excitation of core 2): and third, to reconnect one of the potential leads and

to restore the null by adjusting the core 2 excitation.

The process is repeated until the adjustment of YJY-2

converges to a fixed ratio, which is equal to the open

circuit ratio of the transformer potential windings.

A fast convergence of this series of balances is obtained if the equivalent impedences in series with

the potential windings are very small with respect to the impedences of the second core excitation circuit and of the Wagner circuit. A technique for eliminating

the impedance in series with the high-voltage secondary winding of a three-winding transformer at the ex- pense of slightly increasing the impedence in series

with the low-voltage secondary winding has been

described [6]. This technique is illustrated by the

equivalent circuits based on ideal transformers shown

in figures 10a and 10b. The circuit of figure 10a is equivalent to the circuit of figure 10b. with z\ = Z\ — 9z; , and 2-1 = z-2 + 0.9zx for a transformer with a 10:1 ratio. The unlabeled impedences and turns ratio are not precisely equal in the two figures, but this is not important. The significant result is that 23 can be

adjusted to make z\ =0.

A technique for making the effective impedance in series with the low potential winding also equal to zero

is shown in figure 11. If there is no current in the low

potential winding, there will be no voltage across transformer T>, and hence no current through the

FIGURE 10. Equivalent circuits for showing the effect of z,i on z\ (see text).

auxiliary admittance Y». However, if there is a current

in the potential winding, T> will be excited, producing a compensating current in Y'.,. In the worst case, sup- pose the potential lead of Y-> is shorted. Then \{z'.l = z!,, the voltage on the 100-turn winding of T> will be eJ2.

and the voltage applied to Y'2 will be equal to e\. The

drastic measure of shorting the potential lead of Y> would then cause no deflection of the null detector

provided Y!, = Y->. An argument based on the circuit

linearity indicates that smaller currents in the low

potential lead would likewise cause no deflection of the

null detector.

Figure 11. Compensation scheme for eliminating the effect of z >.

Applying the concepts of figures 10 and 11 to the

basic circuit of figure 9 we have a system in which the currents in both potential leads can be adjusted to zero, but in which the bridge balance condition is not

strongly dependent upon either auxiliary balance. In order to make the potential lead impedances definite so that 23 and z'i need only be adjusted once during

construction of the bridge, we have elected to plug the

star connectors directly into the potential terminals on top of the bridge. The adjustments of 23 and z'[ then serve to make the effective impedances from the

internal junction points of the star connectors to the

equivalent voltage generators of the transformer potential windings equal to zero, using an auxiliary

admittance Y'i = Y-i. It is to be noted that when this

484
bridge was designed, the special defining transformers

described in section 2 had not yet been invented and

the potential leads were open circuited by simply

unplugging them. This technique is adequate in this particular case since the potential lead equivalent

series impedances are very small. When the junctions

between the current and potential leads are located

remotely from the bridge, as is the case when measuring four-pair admittances, the lead impedances are

not small, and the Wagner and second core excitation

adjustments are very critical. The shunt capacitance

uncertainties caused by unplugging the coaxial

connector could then cause substantial errors. In order to make a direct reading ratio set out of the

circuit of figure 9. a means must be provided for adjusting the ratio of the potential winding in accurately divided steps. The adjustment of the magnitude

of the ratio is straightforward, and can be accomplished

in principle by means of a step-down transformer T* and a six-decade inductive voltage divider T3 magnetically coupled to the potential windings of 7\ as shown

in figure 12. Although 7s and are represented as being ordinary single-stage transformers, they are

actually both two-stage devices. The first stage of T3 is coupled to the first stage of T\ only, and is used to excite the first stage of 7Y The switching of the first and second stages of Tz are ganged, which improves

the accuracy of the voltage division, and results in a

negligible interaction with the core 2 excitation

adjustment for 7\.

Constructing an accurate quadrature adjustment for a direct-reading ratio set is much more difficult than

constructing an equally accurate magnitude adjust- ment. At the heart of the problem is the requirement

for a 903 phase-shifting circuit. The conflicting requirements of small output impedance, relatively large

output voltage, and reasonably small power dissipation

tend to limit the accuracy and range of most passive

quadrature balance circuits.

~7h

20T 20T

Irv-J"; 200

T

R2

-00'

FIGURE 12. Circuitry for adjusting the real and quadrature com- ponents of a transformer ratio.

In the quadrature balance system shown in figure 12,

the output of a phase shifting circuit consisting of a

0.01-/xF capacitor d and a 10-O resistor /?, is con- nected to the primary of the three-winding transformer

T6 . One secondary of Ta is in series with the low

potential winding of Tx . R x is small compared to the

input impedance of T*, and since w/?,C t <§ 1, the volt- age eq added to Tx by T6 is nearly orthogonal to the

voltage across Tb and nearly of the desired magnitude.

A test of the voltage added to Tx by T6 can be made by

setting up a bridge using a precision 1-nF capacitor

C2 and a precision 10-fi resistor R». A detector con- nected to terminal D' will register a null when e$

j(oC> = e (,//?>. A small complex adjustment of the

excitation to the primary of T6 may be made to give a

null at D' . so that e? =e^'(uC>/? ». Divider Th is a six- decade, two-stage device similar to T3 , and in fact has

separate windings on ganged switches to separate the coarse output tap connected to Cx from the precision

output tap connected to C2 . Resistor R> may at this stage be considered to include the equivalent series

impedance of the winding on T6 to which it is con- nected, but in the final design an extra winding was put on T6 to provide a four-terminal connection for R>. When the circuits of figures 9, 10. 11, and 12 are combined, the result is a system with two sets of detector terminals. The advantages of combining the

two to give a single detector whose response is essentially independent of the excitation of T6 is by now quite

apparent. The NBS bridge has been designed so that connecting an auxiliary admittance Y'!=Yo from D' to the main null detector D indicated in figure 9 achieves

the desired effect. Rather than using two equal

auxiliary admittances Y!z and F.", a single admittance is used for both purposes by connecting the 200-turn

output winding of T* in figure 11 between terminal D'

in figure 12 and the auxiliary admittance. This serves

to add the compensating voltage produced in the

quadrature balance circuit of figure 12 to that produced

in the load compensation circuit of figure 11. The complete four-pair direct-reading ratio set contains all of the features described above, and some

special compensation circuits for obtaining a quadrature adjustment which is precisely proportional to frequency. The only critical impedances required in the bridge are the capacitor Ct and the resistor /?_> involved in the quadrature adjustment circuit. The bridge was a test bed for a number of previously

untried ideas, and as such it grew in a rather haphazard

manner. It is felt that the bridge could be greatly improved if it were rebuilt, and while this is not contemplated at this time, a complete circuit of the existing

bridge would be more exposing than revealing, and

therefore no such circuit appears in this paper. It

is perhaps useful to point out that the key to achieving

highly accurate and stable ratios in critical trans- formers and inductive dividers is to make generous use

of multi-stage transformers with magnetic shielding

between stages. The principal errors in such trans- formers are due to capacitive loading effects working on their output impedances. Both the interwinding

485
capacitances and the output impedances are minimized

by minimizing the number of turns on the transformer.

The optimum number of turns on a shielded two-stage

transformer designed for audiofrequencies is believed

to be less than 100 turns, which is much smaller than

the number normally employed in a single-stage

design.

The calibration of the bridge was in some ways even more formidable than its construction. The first step was to check the linearity of the bridge dials at several

frequencies by an external calibration technique in which a fixed admittance was repeatedly added in parallel with one side of the bridge for various settings

of an auxiliary admittance in parallel with the other

side. Both magnitude and quadrature balance linearities of all dials were checked in this way. A resolution

of 1 part in 10 10 was obtained at 1592 Hz. and of 1 part

in 109 at all other frequencies.

The second step was to determine the actual bridge

settings which would produce an exact 10:1 ratio. A

modification of the permutation method described in an earlier paper [12] was used for this purpose.

The last step in the calibration was to determine the

actual magnitudes and phase angles corresponding to changes of the real and quadrature dials of the bridge over their entire ranges. A quadrature bridge and a system for comparing admittances in a 1000:1 ratio are needed for this step. Three step-ups using the 10:1

ratio of the bridge under test provided the 1000:1 ratio with ample accuracy.

The bridge accuracy is optimum at 1592 Hz. at which

frequency the ratio is adjustable over a range of ±5

parts in 104 in steps of 1 part in 109 for both real and

quadrature components. A complete calibration using

the procedure outlined above disclosed that the bridge

readings are linear at 1592 Hz over the entire range of

the dials within an estimated uncertainty of 3 parts in 10 10

. The actual changes in both real and imaginary

parts of the bridge ratio produced by changing the

dials over their entire ranges differ slightly from the

values indicated on the dials. The discrepancies are never greater than 5 parts in 10s

. even when the bridge

dials are at the limits of their ranges. Corrections for linear errors of this form are relatively easy to apply.

At 159.2 Hz. the range of the bridge quadrature bal- ance control decreases to it 5 parts in 105

. and at 15920 Hz it increases to ± 5 parts in 10:i

. The first decade of the quadrature balance control is somewhat

nonlinear above 10 kHz, but if the bridge is not used for comparing admittances for which the quadrature component of their ratio exceeds ± 5 parts in 104

, no errors exceeding 5 parts in 109 exist from 159.2 Hz to 15920 Hz. The additional circuitry needed when using the

bridge for comparing four-pair standards. Pi dndP>. is shown in figure 13. The critical leads to the standards

Pi and Pz are terminated at the reference planes of the

defining transformers TV, T», and jT9 , and the leads be- tween these transformers and the standards are

treated as part of the standards. The combining net- work involving 7\o and the associated phase-shifting

network, which joins the detector terminals Di and £)_> of the two standards, is adjusted to obtain a null on D«

when an oscillator is connected to the 100-turn input

winding of the defining transformer TV. Then the input

winding of T* is shorted and the oscillator is recon- nected normally. Alternatively, this adjustment can be

made by leaving the oscillator connected normally, and

adjusting the network so that Dh does not respond when

the short on TH is removed.

FIGURE 13. Comparison of four-pair admittances with the four-pair

DRRS.

The Wagner balance and second core excitation of the four-pair bridge are adjusted to obtain nulls on

detectors connected to points Z)4 and of defining

transformers TV and Ty rather than by disconnecting

the potential leads to the bridge. With this system the

bridge ratio is defined at the reference planes in TV and

7», and differs from the ratio defined at the potential

terminals of the bridge. Techniques for comparing the

ratios defined in these two ways can be easily devel- oped, or alternatively the permutation method for measuring the bridge ratio [12] can be modified to give

the ratio at the reference planes of T- and 7» directly.

A standard pair of cables for connecting Tr and Tit to the potential terminals of the bridge is obviously

required.

The circuit of figure 13 is not easy to operate directly,

partly because the deflections of the auxiliary detectors

connected to DA and both change when either the

bridge Wagner Balance or the second core excitation

is changed, and partly because these balances are both

much more critical than they are when comparing twopair admittances. The criticalness is caused by the extra equivalent impedances in series with the bridge

potential terminals resulting from the added cables

between these terminals and the internal junctions of Pi and Pi. A network with the double purpose of

combining the output of Z)-, with the main detector and

of providing two auxiliary detector outputs, one of which responds to each bridge auxiliary balance, is shown in figure 14. The 100-turn windings of both T7 and 7» are terminated with 50-Q resistors, which when referred to the

486
1-turn input windings yields a very small impedance.

The voltages across the 100-turn windings are thus

proportional to the currents in T- and T». The output

voltage at terminal DH is proportional to the sum of

these two currents, and responds to changes in the

FROM BRIDGE

AUX. OUTPUT \

) ZH— soa

£0T[5OT

o7 LrricU o(

50O

TO Yj I00T

figure 13 for combining several null detectors causes no appreciable reduction in bridge sensitivity when

the admittances under test are capacitors, but when

one is measuring ac resistors this technique substantially increases the thermal agitation noise appearing at the detector. The alternative combining network

shown in figure 16 avoids this problem, and works well when the smaller of the resistors under test is 10 4 Cl or

less. The alternative combining network has been

FiGIRE 14. Combining netuork for assuring fast convergence of DRRS auxiliary balances. Wagner balance but not to changes in the second core

excitation. The output voltage at terminal D- is proportional to the current in T- minus 1/10 of the current in

Ty. It responds to changes in the second core excitation, but not to changes in the Wagner balance. A very

rapid convergence of the auxiliary balances can thus be

obtained.

A complete combining network for joining the two

auxiliary detector points D- and D* with the main

detector DH would require two complex adjustments for a total of four adjustable parameters. The simple

combining network of figure 14 contains only one

adjustable resistor and does not give perfect compensation. It is adequate at 1592 Hz and below because the reactances of the bridge cables are much smaller than

their resistances at this frequency, because the impeddance of the high-voltage potential lead is onlv onetenth as important as the impedance of the low-voltage

potential lead, and because a factor of 10 reduction in the criticalness of the auxiliary balances was found to be sufficient. A detailed calculation to verify the above conclusions

is not appropriate to this paper. The analysis can be

carried out with the aid of the equivalent circuit for the bridge potential circuit shown in figure 15, in which

ii represents the second core excitation, i> represents

the effect of changing the Wagner balance, is and / 4 are the currents in the potential leads, id is the short

circuit current at the main detector, and Y\ and Y2 are

the admittances under test. The voltage generator in the current leads of the bridge is represented by e,, and

can be set equal to zero when analyzing the behavior

of the auxiliary balances.

The use of an auxiliary admittance Yl as indicated in

FiGLRE 15. Equivalent circuit representing the DRRS auxiliary- balances and their effects.

K9 (&

FiGIRE 16. Alternative combining netuork for use when Ym&IO" 4 siemen.

found to be more convenient than the circuit of figure 13 for the measurement of all admittances for which

I Y> | 3= 10~ 4 Siemens. The circuit of figure 13 is usually

easier to adjust when \ Y-> \ =£ 10 -5 Siemens, because it is not as strongly affected by admittances to ground at the junction ofFi and Y> and in the combining network

involving T\ 0. The .circuit of figure 14 works equally

well with either combining network.

Although no coaxial chokes are shown in any of the

circuits for this bridge, they are obviosuiy essential for proper operation. They are inserted where needed to break ground loops, and tested in accordance with the

principles laid out in section 2.

5. Equal-Power 1 00 : 1 Resistance Bridge

The comparison of two resistors whose ratio substantially differs from unity is usually made either by

connecting them directly in series and comparing their voltages, or by applying equal voltages to them and

comparing their currents. As a result, the powers dis- sipated in the two resistors differ greatly, and if the

load coefficients of the two resistors are about equal,

the excitation of the bridge is limited by the load coefficient of the resistor which dissipates the must

power. In this case, the bridge sensitivity is much less

487
than would be obtained with a unity ratio bridge containing resistors of comparable load coefficients. Using a voltage transformer bridge such as the direct- reading ratio set described above, the larger resistor

dissipates ten times as much power as the smaller

resistor. Using a current transformer bridge such as the conjugate bridge obtained by interchanging generators and null detectors, the smaller resistor dissipates ten times as much power as the larger resistor.

If a bridge contains a voltage transformer of ratio N\ and a current transformer of ratio N->, then the bal- ance condition is Ri= NU\>R>; and the powers dis- sipated in the two resistors are equal if Ni =N*. The

bridge sensitivity in this case is equal to that of a unity

ratio bridge for comparing equal resistors. The tech- nique is only applicable for resistors whose ratio is a

perfect square, but this is not a fundamental limitation

since a rational number can be found that is arbitrarily

close to any irrational number.

For the very important case in which the two resis- tors to be compared have a ratio of 100, each trans- former must have a ratio of 10. The bridge described in section 4 above contains all of the components re- quired of the voltage transformer in such a system, and

in principle another bridge exactly like it but with a subtractive rather than an additive ratio could be used

for the current transformer. Fortunately the current transformer part of the bridge need not be provided

with an adjustable ratio, and is therefore much easier

to build. A quite different approach to an equal-power,

four-terminal (but not four-pair) impedance bridge has

been developed by Henry Hall of the General Radio

Company.2 Both two-pair and four-pair equal-power

bridges have been described in the literature [12], but

the operation of the four-pair version is not clear unless

two-stage transformers are employed.

- A paper describing this work was presented at the 1970 Conference on Precision Electromagnetic Measurements at Boulder. Colorado. The text of this paper will be published in an issue of the IEEE Transactions on Instrumentation and Measurement.

The 100:1 bridge used at NBS for comparing a

10 3-D four-pair resistor with a 105 -fi four-pair resistor

at 1592 Hz is shown in figure 17. It is convenient for the purpose of analyzing the behavior of this circuit mentally to interchange all generators and detectors

and to work with the conjugate bridge. Although the

circuit could of course be described directly without

making use of the fictitious interchange, and although

this would seem a more natural approach to people

familiar with current comparators, the interchange is helpful here because it allows the use of the same

terminology as was developed for describing the direct

reading ratio set of section 4.

If we consider then that a generator is connected to terminal B in figure 17, it can be seen that the second

core excitation circuitry is identical with that described

in section 4, but that the Wagner circuitry is much

more complex. The Wagner circuit combines a 10:1 voltage transformer T{5 with a 10:1 current transformer

7\6 to obtain simultaneously nearly the correct 10:1 voltage ratio and 10:1 current ratio in the two resistors

under test. Note that the ratio of currents in the two

windings of Tl6 is nominally equal to the turns ratio, so

that very little voltage appears across either winding.

The two auxiliary adjustments then provide only the

small corrections necessary to meet exactly the re- quired conditions of zero current in the defining

transformers Tv> and Tl3. Although a much simpler Wagner balance system

involving resistors could have been used, the circuit chosen is superior because it leads to a negligible

augmentation of the thermal agitation noise appearing

at a detector connected at point B. This consideration is of no importance in the Wagner circuit of the voltage

transformer part of the bridge, and the use of R3 to extend the limited range of the internal Wagner

adjustment of the direct-reading ratio set is perfectly

satisfactory.

FIGURE 17. A 100:1 equal power bridge making use of the four-pair DRRS.

488
The auxiliary resistor required for proper operation

of the quadrature balance controls of the direct-reading

ratio set would most conveniently consist of a 10 4 -O

resistor connected between the auxiliary output

terminal of the bridge and the detector point B. This

would produce an unacceptably large thermal agitation

noise. The use of a 500-H auxiliary resistor for Y'< which is connected to a tap on Ti$ eliminates this problem. This does not interfere with the use of the

circuit of figure 14 to reduce the criticalness of the

direct-reading ratio set auxiliary balances.

A circuit similar to that of figure 14 and connected

to defining transformers T\> and T\z is used to reduce

the criticalness of the current transformer auxiliary

balances and to provide a means of quickly making the

current transformer auxiliary balances. This circuit is shown in figure 18. An auxiliary admittance Fn (not

shown) is connected between this circuit and the

generator terminal A on the direct-reading ratio set.

FIGURE 18. Combining network for assuring fast convergence of

current transformer auxiliary balances.

The circuit of figure 18 is complimentary to the

circuit of figure 14. In use, the current transformer

auxiliary balances are adjusted so that temporarily

connecting a generator to either G> or G3 produces no change in the detector connected to terminal B in figure 17. The adjustable 50-Q resistor in figure 18 or the auxiliary admittance Y3 is set so that these auxiliary

balances are not critical.

It was necessary to compromise in choosing the

number of turns for transformer 7\ 4 in figure 17 in order to achieve a reasonably stable ratio without

unduly augmenting the thermal agitation noise due to excessive shunt conductance in the transformer. With

200 turns on Tu, the shunt conductance measured at point B with Pi and P-z removed was found to be

1.8 X 10-5 siemens which is only slightly less than the

2 X 10 -5 siemens conductance contributed by the

resistors under test. With 4 x 10 -3 W dissipated in each resistor (20 V on Pi), the sensitivity is sufficient

to resolve 1 part in 10y using a phase-sensitive detector

followed by a recorder, which is considered to be

adequate.

The easiest way to measure the ratio of the current transformer Ti4 is to interchange the connections with

respect to the direct-reading ratio set so that a balance

is obtained when Pi is nominally equal to P>. Inter- changing Pi and Pa and rebalancing allows the current transformer ratio to be determined from the known ratio of the direct-reading ratio set. A slight change in the

Wagner' circuit of the current transformer is also re- quired. This consists of interchanging the windings of

Tin. Resistors cannot be used for Pi and P* because of

their large load coefficients, so a pair of 1-nF capacitors

is used instead. The voltages on the two capacitors

change from 20 V to 200 V in the course of the measurement, so their voltage dependencies must be accurately

known. Techniques for measuring the voltage dependencies of capacitors to the required accuracy have

been described by Shields [14J.

6. Conclusions

The series of bridges described above were built to provide the admittance comparisons necessary for an

absolute ohm determination, as sketched in the intro- duction. It has been found that all of the comparisons

necessary for this work can be made with an overall uncertainty of a few parts in 109 using these bridges. In

practice, the largest source of uncertainty results from

instabilities in the admittance standards used in the

measurements. This problem can be handled only be

carefully controlling the temperature of the standards.

7. References

[1] Cutkosky. R. D.. Four-terminal-pair networks as precision admittance and impedance standards, commun. and Electronics 70, 19-22 (Jan. 1964).

[2] Homan, D. N., Some techniques for measuring small mutual

inductances. J. Res. Nat. Bur. Stand. (U.S.), 70C (Eng. and

Instr.), No. 4, 221-226 (Oct.-Dec. 1966).

[3] Haddad. R. J., A resistor calculable from DC to <o= 103 rad/s. Master's thesis. School of Engineering and Applied Science,

The George Washington University, Washington, D.C., 1-57

(Apr. 1969).

[4] Thompson, A. M., AC bridge methods for the measurement of three-terminal admittances, IEEE Trans. Instr. Meas. IM-13, No. 4. 189-197 (Dec. 19641.

[5] Kavanagh. R. J., Noninteracting controls in linear systems.

Trans. AIEE 76, part 2, 95-100 (May 1957).

[6] Cutkosky, R. D., Active and passive direct reading ratio sets

for the comparison of audio-frequency admittances, J. Res.

Nat. Bur. Stand. (U.S.), 68C (Eng. and Instr.), No. 4, 227-236

(Oct.-Dec. 1964).

[7] Cutkosky, R. D.. Evaluation of the NBS unit of resistance based

on a computable capacitor. J. Res. Nat. Bur. Stand. (U.S.), 65A (Phys. and Chem.), No. 3, 147-158 (May-June 1961).

[8] Thompson. A. M.. An absolute determination of resistance based on a calculable standard of capacitance, Metrologia 4,

No. 1. 1-7 (Jan. 1968).

[9| Millea. A., and Hie. P.. A class of double-balance quadrature

bridges for the intercomparison of three-terminal resistance, inductance and capacitance standards, Metrologia 5, No. 1, 14-20 (Jan. 1969).

[10] Brooks. H. B.. and Holtz. F. C, The two-stage current trans- former. AIEE Trans. 41, 382-393 (1922).

489
[II] Hamon. B. V., A 1-100 fi build-up resistor for the calibration of [14] Shields, J. Q., Voltage dependence of precision air capacitors. standard resistors. J. Sci. Instr. 31, 450-453 (1954). J. Res. Nat. Bur. Stand. (U.S.). 69C (Eng. and Instr.), No. 4.

[12] Cutkosky, R. D.. and Shields, J. Q., The precision measurement 265-274 (Oct.-Dec. 1965). of transformer ratios. IRE Trans. Instr. 1-9, No. 2, 243-250

(1960).

[13] Clark, H. A. M.. and Vanderlyn. P. B.. Double-ratio AC bridges

with inductively-coupled ratio arms, Proc. IEE 96, Pt. 3, 189-202 (1949). (Paper No. 74C3&4-299)

490
A VARACTOR NULL DETECTOR FOR AUDIO FREQUENCY

CAPACITANCE BRIDGES

BY

R.. D. CUTKOSKY

Reprinted jrom IEEE TRANSACTIONS

ON INSTRUMENTATION AND MEASUREMENT

Volume IM-17, Number 4, December, 1968

pp. 232-238 Copyricht © 1968

The Institute of Electrical and Electronics Encineers, Inc. printed in the u.s.a.

491
A Varactor Null Detector for Audio Frequenc y

Capacitance Bridges

ROBERT D. CUTKOSKY, member, ieee

Abstract—A two-varactor, double-sideband up-converter pumped

at 30 MHz has been constructed. Operated at room temperature with

a signal frequency of 10* rad/s, the device has an optimum source

resistance of 10s ohms and a minimum noise figure of 0.01 dB. Im- mersed in liquid N2 , minimum noise figures below 0.001 dB referred

to a room temperature source have been measured. The device is particularly useful as a null detector for audio frequency capacitance

bridges. At 104 rad/s, a signal current of 10~H amperes through a capacitance of 1000 pF can be detected in less than one second with

this instrument. Techniques for suppressing microphonics and other extraneous sources of noise are described.

I. Introduction

BRIDGE circuits commonly used with calculable capacitors may be viewed at their detector ter- minals as a signal current source of 10-14 amperes

at an angular frequency of 10 4 rad/s (1592 Hz) shunted

with a 1000-pF capacitance and a 10_9 -reciprocal-ohm

conductance. Thermal agitation noise is produced by

the conductance, and is given by 2 n = 4X10~ 15 ampere

for a bandwidth of 1 Hz. Since this noise current is considerably smaller than the signal current, detection of the signal would seem to be a simple matter of matching

the source to a detector having a noise figure smaller than a few dB.

Although conventional detectors are available with

noise figures below 1 dB, the capacitive source cannot

be correctly matched to these detectors without greatly

increasing the thermal agitation noise. For example, a 10-henry inductor will correctly tune out the 1000-pF

capacitance at 10 4 rad/s, but even if it has a Q of 500,

it will augment the noise current in a 1-Hz bandwidth

by an amount 2 n ~ 1.8 X 10~ 14 ampere, and in addition, the tuning will require readjustment whenever the source capacitance changes.

If the detector noise is sufficiently small, it is pref- erable to connect the source directly to the detector with no matching network. A detector with an optimum

source impedance of 105 ohms and a noise figure of about 0.0025 dB is called for in this application if a 1-Hz bandwidth is used, and a detector with a noise

figure of 0.01 dB would give a performance comparable

with that obtained using a matching inductor having a Q of 500, but retuning would not be required to compensate for changes in source capacitance.

Manuscript received June 17, 1968. This paper was presented at the 1968 Conference on Precision Electromagnetic Measurements,

Boulder, Colo. The author is with the National Bureau of Standards, Washington, D. C. 20234.

-Varactor amplifiers are extensively used at microwave

frequencies in very low noise detector systems. Audio

frequency varactor amplifiers have not been as extensively investigated, but noise figures below 0.1 dB have

been reported for an amplifier designed for the range

2 to 50 Hz [l ]. An investigation made at NBS indicated that it should be possible to construct audio frequency

varactor amplifiers with noise figures below 0.01 dB.

This paper describes recent progress toward realizing these theoretical predictions.

Although very low noise amplifiers are of critical importance for capacitance bridges, other uses for the sys- tem described here may be found. One possible field of application is the detection of signals from very low

temperature resistive sources.

II. Circuit Description

The circuit investigated in our laboratory contained a pair of varactor diodes in a balanced circuit pumped

at 30 MHz. An audio frequency input signal results in the production of sidebands separated from the 30- MHz carrier by the signal frequency. A commercial

30-MHz amplifier intended for use as an intermediatefrequency amplifier in a superheterodyne receiver (IF

amplifier) amplifies these sidebands, which are then detected in a phase-sensitive detector whose reference voltage is obtained from the 30-MHz oscillator. A dc

component in the output of this detector indicates a lack

of balance in the varactor pair, and is returned as feed- back to the bias circuits to eliminate the carrier from the output. This precaution prevents saturation of the IF

amplifier, and helps to stabilize the overall system.

The audio frequency output of the 30-MHz phasesensitive detector contains the amplified signal at its original frequency, and is usually passed after further amplification through a second phase-sensitive detector whose reference is obtained from the oscillator which

drives the audio frequency bridge. The original signal

has at this point been reduced to direct current, and is easily filtered to reduce the bandwidth of the detector system, thereby reducing the noise. The construction of the system is illustrated in Figs.

1-7. The varactor head, Fig. 2, is contained in an evacuated chamber, Fig. 4. This chamber is immersed in liquid N2 contained in a 4-liter Dewar. The Dewar is enclosed in a rectangular mu-metal box visible in Figs. 6 and 7. The operation of the varactor head, Fig. 2, is as fol- lows. A 30-MHz-pump input drives the two varactors 492
CUTKOSKY: VARACTOR NULL DETECTOR

SIGNAL

INPUT

(IO'°OHMS)

30 MHz PUMP

INPUT

MVI750 3E 2000pF A L MVI7S0

OUTPUT

lOOOpF —)hr)^ lOOOpF CHOKE t 3 CHOKI

820 pF —)H-)( 820pF

6.3 VOLT SUPPLY

82kfl | *k i •«» FRoST/d.

I I j 18 V( 10 TURN, lOkfl TUNING

lOTURN.IOOkfl

Fig. 2. Varactor head and bias supply circuits. Section within dotted lines is maintained at 77 K. Fig. S. Bottom view of varactor head showing L\, Lz, L3 , and Lt .

30 MHz OUTPUT

9kni ?i.5Mn

I6>»F

.01 ^

51—, Ml 1 r-L }r005^F 31

10 TURN OUTPUT

LEVEL

CONTROL

joijiF

33. 3f

MVI750 j[

;ii3on lOOkfl^ iooka

w 5pF

o + ICO VOLTS

^Q82

*0-25pF 1 12 AT7 53*

I 5pF IN3'

INPUT FROM

360fl IF AMPLIFIER W» r O 360 a I

X

LX

AT SIGNAL OUTPUT

jOI^F S270n

10/iH 390pF^

30MHz, 3rd OVERTONE fff rft

Fig. 3. Pump oscillator-phase-sensitive detector circuit. 493

750 /*H

io Mn

FEEDBACK OUTPUT
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1968

Fig. 6. Front view of varactor amplifier.

symmetrically to vary their capacitances at 30 MHz.

Inductor L\ is adjusted to resonate with the parallel

combination of the two varactors, and minimizes the magnitude of the 30-MHz voltage appearing at point A

for a given varactor excitation voltage. Inductor L3 resonates with the 180-pF capacitor C\ connected to it, and as a result the pump line is teminated with the 50-ohm resistor R\. This was done so that variations in the length of the 50-ohm pump line would not affect the varactor excitation voltage. The phase shift in the 30-MHz circuit may then be adjusted by varying the pump line length. The choke L4 and the 30-MHz trap

involving Li suppress the high-frequency voltages that would otherwise appear at the signal input terminals.

If these filters were not present, the behavior of the sys- tem would depend upon the impedances of the signal

input circuit at multiples of 30 MHz.

The bias voltages are adjusted to balance the 30- MHz pump currents passing through the two varactors, and to produce a 30-MHz resonance in the circuit containing L6 and the series combination of the two varactors. Under these conditions, the output voltage induced

in the one-turn coupling loop associated with L6 is zero when the audio frequency input signal is zero, but due

to the reversed polarity of the two varactors, the pres- ence of a signal voltage results in an unbalance in the varactor bridge and in the appearance of an output

voltage. A mechanism for adjusting the coupling loop

position is accessible from outside the Dewar-varactor

head assembly, hence the match between L6 and the IF

amplifier can be optimized with the system in operation.

Fig. 7. Rear view of varactor amplifier.

The signal input and bias circuits are designed to minimize the contributions to the output signal of the thermal agitation noise voltages produced in their resis- tive components. The low-frequency series resistance of the combination Li, L4 , and L2 is less than 1 ohm, and is negligible compared with the measured amplifier equivalent noise resistance i?eq (see Table I). Likewise, the low-frequency equivalent series resistances of the bias

circuits were reduced below 1 ohm above 1 kHz by the

use of high-<2, 2-juF bypass capacitors.

Current noise generators operative at audio frequencies include shot noise in the varactor diodes, and

thermal agitation noise in G\. The varactor leakage cur- rents are below 10-14 ampere at liquid N2 temperature

(77K), and the resulting shot noise is not significant.

However, this source of noise would not have been

negligible at room temperature. The 10~10 -reciprocalohm conductance G\ is needed to maintain the dc volt- age levels in the circuit at their proper values, but is unfortunately a large source of current noise. The measured conductance of this element at 1592 Hz was found

to be 3.1 X10-10 reciprocal ohm, which is close to the measured amplifier equivalent noise conductance of

5.2 X10-10 reciprocal ohm (see Table I). It would be

desirable to maintain G\ at 77K in a new construction.

Much has been published on the theory of varactor

amplifiers [2], [3]. The following first-order analysis

leaves much to be desired in elegance, but has been

found particularly useful in the unusual case of an un- matched signal source. A varactor pumped at an angular frequency wP ex494
CUTKOSKY: VARACTOR NULL DETECTOR

TABLE I Experimentally Determined Amplifier Noise Parameters

Based on the Equivalent Circuit of Fig. 9

Frequency

(Hz) C-i) (reciprocal 9.) (reciprocal P.) (dB)

90 400 1.1X10-'° 5.2X10-' 1 .8 X10"3 160 180 4.7X10-'° 1 .6X10"' 2 .5 X10-J 400 100 4.0X10-10 2.0X10-' 1 7 X10"3 1000 35 5.1X10-10 3.8X10-' 1 .2 X10"3 1592 23 5.2X10-'° 4.8X10-' 0 95X10-3 4000 14 2.3X10"9 1.3X10-' 1 6 X10~3 10000 14 1.2X10-8 2.9X10-' 3 6 X10~3 16000 16 2.7X10-* 4.1X10"5 5 7 X10~3 40000 50 6.5X10-8 3.6X10"' 16 xio-3 100000 140 6 X10-7 6.5X10-' 80 X10"3

T

(a)

1 F AMP.

Fig. 8. Varactor equivalent circuits at output frequency, (a) An

individual varactor. (b) Complete output circuit, in which 2 eout = —2 e,(AC/C0 ) sinup/ (see text).

hibits a time-dependent capacitance C=C0 +AC sin

Up/, where Co is the average capacitance, and AC is the peak capacitance excursion from C0 and depends upon

the pump voltage.

If the varactor is open-circuited at the output fre- quency, we have

d

i = — (Cevar) = 0, dt where evar is the varactor voltage. If follows that CevhI = constant = Coe„ where e, is the signal voltage, and

hence

/ AC . \ Co ^1 + sin up

t J

ev«r = C0 e„

and

AC

e, — e, sin a> pt, AC Co

1 H sin uj

Co

or AC — e, sin wp

l, Co

(1)

where eou t is the high-frequency part of eTar . This ex- pression is strictly true only if e, is a constant, but is also valid when e, is a slowly varying signal voltage.

The varactor equivalent circuit at the output fre- quency may then be represented as in Fig. 8(a), where

Rd is the varactor series resistance. Because of the re- versed polarity of the two varactors, their output volt- ages add, and the complete output resonance circuit takes the form of Fig. 8(b), where RT = 2RD +RL , and

Rl includes the series resistances of Lh and the 1000- and 2000-pF bypass capacitors shown in Fig. 2. Thermal agitation noise e„ in Rt is given by e n

- = 4:kTxRTAf where Tti~1"l K. To this must be added

the IF amplifier noise, which under conditions of optimum matching has the effect from the noise standpoint

of adding a resistance i? amp at a temperature Tstd = 290

K

to the circuit of Fig. 8(b). i? amp is given by F= l+i?amp

,'Rt, where F is the amplifier noise figure, or R^mp =Rr

(F—l). The total noise voltage in the circuit is then

given by e n

- (total) ^^kTsR^f+^kT,^ (F-l)AfRT. Using (1), it can be seen that the total noise is equivalent to that which would be produced by thermal agitation noise in a resistor i?eq at rstd = 2°0K at the signal

input terminals given by

C

r ( AC V 1 46r.td.ReqA/- lim I (2 sinwp/J dt —

r-tm J o \ Co / T = ikTNRT Af + 4kT.tdRT (F - 1)A/,

or

(r.td + F Q

(2)

For the IF amplifier used here, F=l.26 (1 dB), and

since Tf//Tltd ^0.26 we have

Rea =

0.26RT /ACcy

\"Co"

A measurement of the Q of the resonant circuit yielded

Rt = 0A ohm. AC/ Co was estimated from the varactor

characteristics and the pump voltage to be AC/Co = 0.1, yielding 7?eq = 10 ohms, in reasonable agreement with the measured results listed in Table I. -Several observations are in order regarding (2). First, the strong dependence upon AC/Co suggests that an

effort should be made to make this ratio as large as

possible. In practice, when AC/C0 is much larger than

0.1 the second- and higher-order terms which were

neglected in the analysis above become very important,

and (2) requires modification. In fact, the amplifier described here oscillates when AC/C0 >0.14.

Second, Rt enters the expression directly, and should

be made as small as possible. Part of Rt is due to the varactor resistances, and is a prime consideration in

495
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1968

their selection. The remaining part is due to the series resistances of L6 and of the bypass capacitors, and this leads indirectly to the choice of a high pump frequency,

since a smaller number of turns are then required for Z, 5 . The resistance RT is also affected by temperature, and

this was a factor in deciding to operate at 77K.

Third, since Tt\/Tst d ~ F — 1, no more than a factor of two can be gained by either reducing the varactor temperature assuming RT is constant, or reducing the IF amplifier noise figure; a substantial reduction in 7?eq requires reducing both terms. Fourth, neither the varactor capacitance nor the pump

frequency enters directly into (2). This is not inconsis- tent with the Manley-Rowe equations, which govern

the varactor power relationships. The effect of increas- ing the pump frequency is to allow the reduction of the

series resistance of Z. 5 , to provide an increased bandwidth Aw, and to reduce the Q of the resonant circuit, which simplifies tuning. We have for the circuit of Fig. 8 (b),

Aoj 1 Co — = oipRr —

Q 2

or, neglecting RL , Aw 1 = — = oj pRdCo,

«p Q

so that

Aw = — = RdC0oj p

~ and Q = • Q oi pRdCo

For the amplifier described here, /P = 3X107 Hz and

Q = 250, so Af=fp/Q » 105 Hz. The tuned circuit should

therefore pass both sidebands generated by signals with

frequencies up to 50 kHz.

In practice, the advantages of using a pump frequency

appreciably higher than 30 MHz would be partially off- set by the larger noise figures that are usually associated

with amplifiers designed for these frequencies.

When the varactor head was first operated in liquid

N2, a very large current noise was observed. This was

traced to a microphonic sensitivity stimulated by the

boiling nitrogen. The microphonic effect was principally due to the use of a polytetrafluoroethylene (PTFE)

insulated wire for the signal lead entering the varactor

head. Static charges on the PTFE induced currents into the signal lead when the lead moved with respect to the ground potential tube in which it was contained. The

wire was replaced with an uninsulated length of thin- wall stainless steel tubing securely anchored at both

ends, and the remaining PTFE insulators in the var- actor head were deionized by passing a flame quickly over their surfaces. Although a rather large piece of PTFE was used to support inductor Ls (see Fig. 4), it is in rigid contact with the inductor, and is a considerable

distance from other ground potential objects. The deionization procedure and the reconstruction of the input

lead seem to have eliminated the microphonic effects. The 30-MHz pump oscillator-phase-sensitive detector circuit shown in Fig. 3 is conventional and needs no

discussion. The filament voltages for it and for the IF

amplifier are obtained from a 6.3-volt negative dc supply

to minimize interference from power line harmonics.

This supply is also used to power the IF amplifier gain

control circuit. The amplifier system requires about two hours to cool from room temperature to 7 7K, and was found to be

quite stable and easy to adjust. Only the tuning of the

circuit involving Ls and the varactors is critical, and

this is done in a way which minimizes the equivalent

noise resistance 7?eq at 1592 Hz.

III. Measured Noise Parameters

Most of the experimental work on this amplifier was

done with a signal frequency of 1592 Hz. Several measurements were made of the correlation capacitance at

this frequency using a 1000-pF source admittance (see Appendix), and it was found to be about 300 pF, which was equal to the measured input capacitance of the

amplifier. This is the expected result based on the analysis in Section II of this paper. At all other frequencies

the above result was assumed to be true, but was not

verified. No measurements of the correlation conductance were made, and it is assumed to be zero. All noise measurements were made at the selected spot frequencies by the use of tuned postamplifiers having bandwidths of a few hertz, and with a thermocoupledc voltmeter combination. The postamplifier noise was

small in most cases, but was not subtracted from the quoted results. Considerable interference from power line harmonics was observed below 1 kHz. A narrower passband postamplifier would have avoided this problem, but the thermocouple response would then have been more diffi- cult to interpret. Better magnetic shielding of the varactor head would probably reduce this interference. Table I lists the pertinent noise parameters for the

amplifier, using the equivalent circuit of Fig. 9. An attempt was made to measure i?eq at 400 kHz. Al- though the results were not conclusive, i?eq appeared to be about 1000 ohms. This result and the data in Table I are consistent with the 50-kHz passband estimation

made in Section II of this paper. The dependence of Geq upon frequency is partially

attributable to the increased ac conductances of G\, C\,

and the associated leads inside the varactor head at higher frequencies. A reduction in Ceq would therefore

result if these conductances were reduced, or if the temperature of Cm were reduced. i?eq increases as ex- pected at high frequency, due to the bandwidth limitation of the 30-MHz tuned circuit. The behavior of i?eq

496
CUTKOSKY: VARACTOR NULL DETECTOR

<5H NOISE FREE

AMPLIFIER

Fig. 9. Equivalent circuit of amplifier as dediced from measure- ments made at 1592 Hz. C=305 pF, u- = ikTGe ,^f, and en

2 = ikTRtqAf, with Ge<l and given in Table I.

at low frequencies seems to have the character of flicker noise (i?eq « 1//), but power line interference may have

given abnormally high noise readings.

When this amplifier was initially operated at room

temperature, a minimum noise figure of 0.01 dB was

measured at 1592 Hz. Unfortunately, after operation

at 77K, the varactor leakage currents at room temperature were very much larger than before, resulting in an

unacceptable level of shot noise at room temperature.

Appendix

A noisy amplifier may be represented by a number of equivalent circuits, including those shown in Fig. 10, where the e's and z's represent current and voltage noise generators.

An arbitrary amplifier may be represented by Fig.

10(a) only if a correlation between the current and volt- age noise generators is postulated [4]. The noise current

ia may then be separated into an uncorrected part ia

' and a correlated part ia —ia

' = Ya 'ea , where YJ is the (complex)correlation coefficient. Fig. 10(b) shows a cir- cuit by means of which an arbitrary noisy amplifier may

be represented in terms of uncorrelated current and

voltage noise generators. An inspection shows the corre- spondence between Figs. 10(a) and 10(b) to be given by:

h = it!

e b = ea Yb = Ya

' (correlation admittance)

Yb + Y„' = Yr The significance of this breakdown is that the correla- tion coefficient needed with Fig. 10(a) appears in Fig.

10(b) as a real admittance, and its effect may be readily

analyzed. In particular, it is clear from Fig. 10(b) that shunt capacitance in the input cable of the amplifier

adds directly to Yb, and hence that the input terminals

to which the circuit parameters are referred must be

carefully specified, even for audio frequency amplifiers.

This physical interpretation of the correlation admittance Yb makes Fig. 10(b) especially useful for representing high-impedance amplifiers.

Fig. 10(c) is the dual of Fig. 10(b). It too uses uncorrelated current and voltage noise generators, yet can

represent an arbitrary amplifier. Its use would be indi- cated for describing low-impedance amplifiers.

Fig. 11 shows the circuit of Fig. 10(b) connected to a

source consisting of an admittance Y. = G,+jB. shunted

NOISE FREE

AMPLIFIER

(a)

(b)

o—vw (^}

Fig. 10. Three possible representations for a noisy amplifier.

NOISE FREE

AMPLIFIER

' v„ ri

:

Fig. 11. A noisy amplifier connected to a source, with source noise

ij2 = ikTGAf.

with a signal current generator i,. Thermal agitation

noise in the source is given by ij- — 4kTa G.,Af, where T,

will be assumed to be standard temperature, 290K.

Since all of the generators are uncorrelated, we have

| Y, + Yb + Yb'Y (3) W\ Yb + Y.\> + i,* + i/- + i b -],

where A is the voltage amplification factor of the sys- tem. Representing the voltage noise by an equivalent

noise resistance i?eq given by eb 2 = ikTR^Af and the current noise by an equivalent noise conductance Geq given by 4

2 = 4kTGe<} Af, we have

£out Y, + Yb + Yh'y- (4)

{i,2 + ikTAfiR^l Yb + K.|« + G. + Gj}.

The noise figure of the system is given by

G. + R^l Yb + r.l'+'c.

F =

G.

Letting Ya = G3 +jB, and Yh = Gb +jBb and differentiat- ing with respect to G, and Bs , we find F is minimum

when B, = —Bb = B°pt

, and

+ Gb

- = G.°pt

.

497
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1968

Lnder the special condition that Gb = Q, we have the in- teresting results:

a:

and

1 4" 2 V Geq i?e

If the term on the right is much smaller than 1, Fm i n

', the minimum noise figure in decibels is obtained from a first- order expansion of the logarithm of Fm , n : Fmin' m 8.68VC7eq 7? cq (dB).

When ]"., is greatly different from I's ° pt

. and in particular

when Bs + Bb y>Gs +Gb, the description of the circuit in terms of noise figure is very awkward, and a direct refer- ence to (3) or (4) leads to much simpler results. In the

latter case, (4) reduces to <W2 « is

2 + IkTAfiR^iB, + Bb )

n

- + G„ + G0 (5) where AkT= 1.6X 10-20 W/Hz, and the smallest signal i, detectable with a bandwidth A/ may be directly inferred from measurements of i?eq , Geq , Bb , B„ and Gs . The following method for measuring the noise parameters of an amplifier is specifically directed toward use with the exceptionally low-noise amplifier described in the body of this paper, for which the approximations

made introduce no appreciable errors. The method is not

likely to be useful at high frequencies, or with very- noisy amplifiers. Since the input admittance of the

amplifier Yin = Yb + Yb

' was found to be almost purely

capacitive throughout the useful frequency range, the analysis will be restricted to the case in which Yb + Yb =jB\a . An instrument is assumed available for measuring relative values of eou t 2 over a selected constant fre- quency band. Let e0 ut-\sc be the value of eont

2 with Ys = <*> (short

circuit). Then from (4), eou t

2

| sc = A 2 -4kTAf i?eq . If a source resistance R,= l/G, is inserted such that eoul

- = 2eout

2

\ sc and if Yb <£G„ F!n «G„ and Geq «Gs ,

eoa :- = A--4kTAf(R^ + R,) = 2A---ikTAfR,q , or i?,.q = R.. The equivalent noise resistance is then simply the value

of that source resistance which makes euut

2 double its sho t circuit value. let eoul -\ oc be the value of eoat - with Ys = 0 (open

cir< uit). Then

eout

2 |oc = A°--lkTAf

so that

Bin

- +

B [n -J

font Gb

- + Bb

°- +

Bin

- « -Vi- (6)

Let £ou t

2

| b be the value of eou t 2 with a selected, measured source admittance Y,' = Gs

' +jB,', with

Then

enul -\ B = A--4kTAf Re Gb

- + (Bb + B.y-

(B in + B/y

+

g: + g,

and

Gb

'- + (Bb + B3 y- +

tout sc (5,„ + B/y

Combining (6) and (7), we have

(Bin + B

(7)

(sin + B,'yN, - BinWx - (B.'y -

Bb

G/

Rea IBJ

(8)

In principle, (8) allows the determination of Bb from

readily measurable parameters. In practice, the admittance Y/ must be carefully chosen to minimize the measurement uncertainties. A technique for measuring Gb can be similarly de- veloped with the use of a source admittance Ye " = Gs " +jB8 ", with |B/'|«GS ". Unfortunately, the source temperature would have to be very low indeed to prevent

its thermal agitation noise from completely masking the

amplifier noise. After measuring Gb and Bb as outlined above, Geq may

be obtained from (6).References

[1] J. R. Biard, "Low-frequency reactance amplifier," Proc. IEEE,

vol. 5, pp. 298-303, February 1963.

[2] H. E. Rowe, "Some general properties of nonlinear elements. II. Small signal theory," Proc. IRE, vol. 46, pp. 850-860, May 19S8.

[3] P. Penfield and R. P. Rafuse, Varactor Applications. Cambridge,

Mass.: M.I.T. Press, 1962.

[4] IRE Subcommittee 7.9 on Noise, H. A. Haus, Chairman, "Representation of noise in linear twoports," Proc. IRE, vol. 48, pp. 69-74, January 1960.

498
ISA TRANSACTIONS: 3 (238-242) 1964

Comparison Calibration

of Inductive Voltage

Dividers*

RAYMOND V. LISLE

RCA Service Company

Missile Test Project Patrick Air Force Base, Florida and

THOMAS L. ZAPFI

Radio Standards Laboratory

National Bureau of Standards

Boulder, Colorado

y Calibrated inductive voltage dividers serve as excellent standards of voltage ratio for use in the comparison calibration of other AC voltage dividers. Differences in phase

angles in the dividers can be accommodated by the production of a voltage in the detector circuit in quadrature with the reference voltage. This paper discusses the comparison circuit, the handling of stray impedances, guarding, measurement procedures, quadrature voltage production, and errors. The comparison method described

can be used in standardizing laboratories for the calibration of inductive voltage

dividers and other voltage ratio equipment.

INTRODUCTION

Inductive voltage dividers may be described as tapped inductors wound on high-permeability cores. The relatively recent metallurgical development of

ferromagnetic core material having extremely high

magnetic permeability has made possible the construction of inductive voltage dividers having excellent

stability and loading characteristics. The effects of

internal loading are inherently small in such inductive voltage dividers, and errors in voltage ratios are frequently less than 1 x 10-6 in the low audiofrequency range. The absolute measurement of voltage

ratios with uncertainties less than this is difficult. However, known voltage ratios embodied in an inductive voltage divider may be preserved as a standard and

re-established again and again. The accurate initial establishment of voltage ratio for the calibration of

inductive voltage dividers* 1-4 ' is not discussed here, but

a method of calibrating an inductive voltage divider by

comparing it with a standard is presented. The comparison method, essentially as described herein, has been

used at the National Bureau of Standards for several

•Presented at the 1 8th Annual ISA Conference and Exhibit, September 9-12, 1963, Chicago, Illinois. tStandards Engineer.

tPhysicist, Radio Standards Engineering Division; Member—ISA

Denver Section.

ISA Transactions • Vol. 3, No. 3

years and appears worthy of general use in standardizing

laboratories. This method provides efficient, accurate

calibration of inductive voltage dividers using common

laboratory equipment with a standard inductive voltage

divider. Comparison circuits have been mentioned in several publications*4-7 ' without detailed elaboration. This paper discusses in considerable detail a particularly successful type of comparison circuit.

THE COMPARISON CIRCUIT

A simple bridge circuit for comparing inductive voltage dividers is shown in Figure 1. The standard

inductive voltage divider is the divider for which the corrections are known. The unknown inductive voltage

divider is the divider for which the corrections are to be

obtained. Other elements of the circuit are the guard divider and shielding circuit, flexible coaxial cables to the detector transformer, the detector, a resistor and

capacitor for the introduction of a quadrature voltage

in the detector circuit, and several fuses. The dashed

lines indicate the connections of a capacitor that may

be used to reduce loading.

The input terminals of the unknown and standard

dividers are joined together by short, heavy leads having

low impedance to reduce voltage drops between the

499
GUARD STANDARD UNKNOWN

r J ' L

-i

ih

Hihi

in

Figure I. Bridge circuit for the

calibration of an inductive voltage divider by comparison

with a standard.

dividers. The electrical connection to the case of the standard divider should duplicate the connection used when the standard divider was calibrated. Similarly, the

electrical connection to the case of the unknown divider should duplicate the connection to be used in subsequent

work. In Figure 1 the cases are shown connected to the common terminals. These connections fix, in a re- producible manner, the stray capacitances from the windings to the case of each divider. Connecting the case to the output terminal is not recommended, as

this would not fix the stray impedances from the case (and output terminal) to the surroundings, which are often quite uncertain and variable. The output of the guard divider is connected to the shield around the detector circuit. When comparing an unknown inductive divider with the standard, the guard divider is set at the same setting as the other dividers and drives the

shield at nearly the same instantaneous potential as that of the conductors within. The guard divider may

be either inductive or resistive because its only function

is to maintain the proper potential on the shield. The

guard circuit minimizes current leakage through the

stray impedances of the detector circuit. Usually there is a relatively large stray capacitance

to ground from the detector transformer shield and the capacitor case. The current in this stray capacitance to ground must not be allowed in the shield of the flexible coaxial cables going to the output terminals of the dividers because a current in the shield can induce a voltage in the inner conductor. To minimize this error, the connection between the detector transformer shield and the capacitor case should be made as short as

possible, and the output of the guard divider should be connected to the shield at some point in this vicinity.

The major components of stray capacitive current then

are conducted directly to the guard divider. The

deleterious effect that is avoided by this procedure may

be observed by increasing the length of the leads in the detector circuit and connecting the lead from the guard

divider to a different point on the shield. If all leads are short the error may be insignificant.

It should be noted that, with the circuit shown in Figure 1, the guard circuit and the capacitor case are

at a potential somewhat above ground, and care should

be exercised in touching the shields to avoid hazardous

shocks. Alternatively, the circuit may be grounded at the output of the guard divider, but the cases of the

dividers will then be at a high potential. With this connection it may be necessary to use an isolation transformer between the power supply and the bridge

circuit to avoid double grounding. Regardless of the grounding system selected, it is good practice to provide an outer covering of insulation for the shielding

wherever needed to prevent accidental short circuits. Many inductive dividers are provided with two "common" terminals connected together and to one

extremity of the winding by a wire of low impedance so

that one terminal may be used with the input circuit and the other may be used to make connections to the

load. In the circuit shown in Figure 1 the inductive

dividers are treated as three-terminal dividers, i.e., the common output terminal is ignored. A small voltage

drop in the internal connection between the two common

terminals would make the three-terminal measurements

differ from four-terminal measurements in which both common terminals are used. However, the difference is usually small and often insignificant. Fuses are inserted

in the circuit as shown in Figure 1 to prevent damage

to components in the event of inadvertent short circuits. The fuse in the detector circuit prevents damage to the

resistor, R, if a large unbalance of the bridge circuit is encountered.

VOLTAGE RATIO MEASUREMENT

Let D + d represent the dial reading of an inductive voltage divider, where D is the reading of the decades to be calibrated (usually the highest three decades) and

d is the reading of the lower decades. In most calibrations D will be a step setting of one of the highest three decades with the other two set to zero. The true ratio, A, can then be represented by D + d + c, where c is the

ISA Transactions July J 964

500
TABLE I

Voltage ratio

d, Cs dz Cx

x 10"6 x 10-' x 10-6 x 10" 0.900 0.0 + 0.4 + 0.1 + 0.3 0.500 0.0 0.0 + 0.4 -0.4

0.100 0.0 -0.1 +0.1 -0.2

0.090 + 0.1 -0.5 0.0 -0.4

0.050 0.0 -0.3 0.0 -0.3

0.010 0.0 0.0 0.0 0.0 0.009 0.0 0.0 0.0 0.0 0.005 + 0.1 -0.1 0.0 0.0 0.001 0.0 0.0 0.0 0.0

Phase angle

Ratiox ys . ^rad Ay, ^rad yz, prad

0.900 000 3 _ 2 0 _ 2 0.499 999 6 + 4 + 1 + 5 0.099 999 8 + 49 + 3 + 52

0.089 999 6 + 36 + 4 + 40

0.049 999 7 + 42 + 4 + 46

0.010 000 0 + 70 + 20 + 90

0.009 000 0 + 60 + 10 + 70

0.005 000 0 + 80 + 30 + 1 10 0.001 000 0 + 330 + 130 + 460

correction to the setting D. For all practical purposes c

is independent of d. At balance Dz + dx + cx = Ds + ds + cs (1) where the subscripts x and s denote the unknown and

the standard, respectively. Solving for the correction

to the unknown divider and realizing that Dx = Ds , one obtains

cx = cs + ds - dx (2)

Ordinarily, dx would be set to zero. Under this condition

cx = cs + ds (3) However, if cx < cs and if the standard divider cannot

be set to a negative ds , the lower decades of the unknown

divider, dx, can be used to obtain the balance. If ds is set to zero the ratio balance equation is

cx = cs - dx (4) Typical data and results for a calibration at 100 V and

1000 cps are shown in Table I.

PHASE-ANGLE MEASUREMENT

A true voltage null at the detector can be obtained if a quadrature voltage is introduced into the circuit to compensate for the phase-angle difference between

(a) (b)

ISA Transactions • Vol. 5, No. 3

the unknown and the standard inductive voltage

dividers. The RC combination in Figure 1 is used to develop and introduce a voltage in quadrature with the output of one of the dividers. If the capacitive reactance of C is made considerably larger than the resistance, R,

the current in R will lead the voltage across the im- pedance R — jXc by nearly n/2 radians and will cause

the quadrature voltage to appear across R. At balance

the current in the detector is zero, and the voltage at the junction, of R and C is Es , as indicated in Figure 2a. An expression for the current in R and C can be written

i = Es/Xe (5) The quadrature voltage, e, is the product of i times R,

e = iR (6) and

e = EsRIXc = EscoRC (7) From Figure 2a, ys and yx are the phase angles

associated with the standard and the unknown inductive voltage dividers, respectively, and Ay is the phase-angle

difference between the unknown and the standard

dividers. An expression for Ay can be written from

inspection of Figure 2a as tanAy = e/£5 (8)

ERROR

Figure 2. Phasor diagram:

(a) voltage phasors; (b) voltage

ratio phasors; (c) phasor dia- gram showing small error from quadrature circuit.

(C)

501
Combining the last two equations yields

tan Ay = wRC (9) For small angles tan Ay x Ay, and

Ay ^ cjRC (10) This equation provides a design formula for the RC

combination in the detector circuit. For any test frequency w a variable capacitor can be chosen in combination with a resistor so that Ay in microradians

is a multiple-of-ten times the reading of the variable capacitor in picofarads. Various multiplying factors can be obtained using resistors differing by multiples

of ten. As an example, at a frequency of 1000 cps,

resistors having values of 1592, 159.2, and 15.92 Q, will produce multipliers of 10. 1, and 0.1, respectively.

Thus a 0-100 pF variable air capacitor can be used to cover a wide range of phase-angle differences. In the above discussion with reference to Figure 2a,

it has been assumed that the phase angle of the unknown

inductive voltage divider is larger than that of the stan- dard inductive voltage divider. Under these conditions,

Ay is positive, and the phase angle of the unknown

inductive divider can be obtained from the equation

Yx = ys + Ay (11)

If the difference yx — ys is negative, the comparison

circuit can be balanced by putting the RC combination on the other side of the detector with R connected to the standard divider. Under these conditions, the phaseangle balance equation would be

yx = ys - Ay (12) Typical data and results from phase-angle measurements

are shown in Table I.

DISCUSSION OF ERRORS

If errors from incorrect connections, incorrect frequency, or incorrect applied voltage (i.e., errors

arising from conditions different from those under which

the standard was measured) are precluded, the ponderable systematic errors, such as those arising from the assumptions made in the derivation of the equations, are of very small magnitude. These will now be

investigated.

The error introduced by the approximation that tan Ay « Ay can be evaluated from the trigonometric

expansion of the tangent, namely,

Ay3 Ay5 tan Ay = Ay + + + . . . (13) 3 15 The second term, Ay3/3, evaluated for Ay = 300 /urad would be of the order of 1 x 10 -11 rad, which is negligible, and higher order terms are smaller yet.

It is convenient to consider a phasor diagram of

voltage-ratio phasors, as shown in Figure 2b, rather than one formed of voltage phasors, as in Figure 2a. The diagram in Figure 2b is formed by normalizing the voltage phasors by dividing the circuit voltages by the input voltage magnitude. The voltage ratio A of an

inductive voltage divider is then £o/|£in|, and the quadrature component, q = e/|£i n |. A small error in ratio from the voltage drop across R

is illustrated in Figure 2c in which the angles have been

grossly exaggerated to illustrate the error. In this diagram A* is the apparent ratio at balance, smaller than the true ratio A' s because the phasor ^ is added at right angles to the phasor As by means of the RC

circuit. The error, A's — As , can be evaluated from the trigonometric series expansion for cos Ay, namely,

Ay2 Ay^

COS Ay = 1 + ... (14)

2! 4! From Figure 2c,

As = ^ cos Ay A x {\ - (15)

The error in ratio is represented by the term A x(Ay2/2\). In comparing two good-quality inductive voltage

dividers at a frequency of 1000 cps a Ay of 100/nrad

would be experienced only at the lower values of ratio below about A = 0.01. The error in ratio would be

approximately 5 x 10-11

. An error of this magnitude

would be negligible in most cases. Another source of error in both magnitude of voltage

ratio and phase angle is the loading on the unknown

divider by the capacitor C. To determine the magnitude

of these errors in practical work, the difference in voltage ratio and phase angle can be observed when

another capacitor having the same value as C is con- nected from the output terminal of the unknown divider

to ground in the circuit in Figure 1 (i.e., across R and

C). The observed differences may be subtracted from

the original readings to yield the values of ratio and

phase angle with no load. Alternatively, the loading

may be eliminated by adding a capacitor as shown by

the dashed lines in Figure 1 : C and C form a capacitive divider to reduce the loading of C on the output circuit of the inductive divider. The magnitude of C must be

adjusted depending on the ratio being measured and,

in particular, must have a value equal to CAs /(\ — As). The power source should have a reasonably low

harmonic content, and the null detector should be

tunable to suppress harmonics in the quadrature

voltage, which would otherwise prevent the attainment

of a precise null. To understand the need for a tunable

null detector, consider that harmonic components

exist in both £s and Ex having a frequency nw, equal

amplitude hEs , and the same phase. The total voltage

across R is

e = eljt + entll (phasor sum) (16) where e,„ is the fundamental component and e,il0 is the harmonic component as shown in Figure 3. The phasor

e„ u rotates n times faster than ew , and its magnitude can

be expressed in terms of the phase-angle difference of the fundamental output voltages, Ay. The expression

is similar to that for equation (7), except the harmonic

current is liEs /Xc and the angular frequency is na>; thus

£>«„ = liEsnwRC = «/i£s Ay (17)

ISA Transactions • July 1964

502
Figure 3. Phasor diagram showing effect of a harmonic in the quadrature voltage.

If this component is not suppressed by the detector, it would not by itself create a systematic error, but it would result in a limitation on the attainment of a null equivalent to a voltage-ratio uncertainty

Prim €nui ?nto uA =_ = —_ = A— = Anhly (18)

|£in| EsiA Es and a phase-angle uncertainty

uy = = nh\y [radians] (19) Es In a typical measurement at 1000 cps, where A = 0.1, n = 3, h = 0.01, and Ay = 20 fxrad, it is found that ua — 0.06 x 10~6 and uy = 0.6 firad. It is evident that a tuned indicating amplifier that suppresses the third harmonic by a factor of 10 would be sufficient to permit

the discrimination of ratio to within ± 0.01 x 10-6 in this case.

It should be noted that if a detector is used that has a

linear response to small signals, it is possible to use it to interpolate between steps on the lowest decade of the

divider.

MEASUREMENT VERIFICATION

A simple partial check on the bridge circuit and the

calibration results can be performed by reversing the

leads to the unknown divider and measuring the complement of the ratio. <4) Caution should be exercised because the case of the unknown divider will now be at a high potential if it has been connected to the common

terminal. It can be seen that 1 — A x is now to be

compared with A.s

. With this connection cx = —(cs + cL + dx). The correction, cx, obtained in this way may be compared with that obtained in the normal

way.

The corrections to both inductive voltage dividers for

the setting of A = 0.5 can be obtained from the complementary ratio measurements without prior knowledge

of either of these corrections. If d6 is the reading of the standard for the normal circuit as shown in Figure 1, and d's is the reading of the standard with the unknown

divider connections reversed, then

cs = -(ds + d's),2 and cx = (ds - d's ) 2 (20) By a somewhat analogous method a check on the phase-angle measurements can be made. The formula

for phase angle can be derived simply from the phasor

diagrams of voltage ratio for the reversed connection.

The phase angles of both inductive voltage dividers set at A = 0.5 can be obtained by the method of complementary measurements without prior knowledge of

either of these phase angles.

CONCLUSION

If care is exercised in setting up the circuit, using good

shielding practices, and a detector having adequate

amplification and harmonic suppression is employed,

the voltage ratios and phase angles of two good-quality

inductive voltage dividers may be compared with

negligible loss in accuracy. Several systematic errors in the circuit, for which corrections could be made, have

been discussed and found to be negligible under the

typical conditions described. The precision of the comparison is limited primarily by the resolution and

stability of the standard and unknown dividers. The

existence of a method for accurate comparison makes

possible the use of calibrated inductive voltage dividers

as standards for the calibration of other dividers and

these, ultimately, for the very accurate measurement of

voltage ratio and impedance ratio.

REFERENCES

1. Sze, W. C, Trans. A.I.E.E., Part 1 76, 444, 1957.

2. Harris, F. K., and W. C. Sze (NBS Washington), unpublished communications to the authors.

3. Zapf, T. L., J. Res. NBS 66C, 25, 1962.

4. Engineering Bulletin No. 1 (Ratio Tran), Gertsch Products

Inc., Los Angeles, California.

5. Engineering Bulletin No. 4 (Ratio Tran), Gertsch Products

Inc., Los Angeles, California.

6. Morrison, N. E., Engineering Bulletin No. 20, Electro

Scientific Industries, Portland, Oregon.

7. Morgan, M. L., Engineering Bulletin No. 29, Electro

Scientific Industries, Portland, Oregon.

ISA Transactions Vol. 3, No. 3

503
JOURNAL OF RESEARCH of the National Bureau of Standards—C. Engineering and Instrumentation

Vol. 68C, No. 4, October-December 1964

Designs for Temperature and Temperature Gradient Compensated Capacitors Smaller Than Ten Picofarads

Robert D. Cutkosky

(July 22, 1964)

A theoretical study is made of the dependence of capacitance upon electrode temperature

in an air dielectric capacitor. The possibility of constructing a {hree-terminal standard whose capacitance depends only upon the temperature of one electrode is pointed out, and some practical capacitors utilizing the principle are described. The temperature sensitive electrodes of these capacitors are constructed of fused silica, which results in temperature

coefficients of capacitance near 0.5 ppm/°C. The temperature independent electrodes of these capacitors may be constructed of any stable metal, facilitating adjustment to nominal

value.

1. Introduction

One of the problems encountered in the construction of standard air dielectric capacitors is that of

reducing the dependence of capacitance upon temperature. The best technique seems to be to construct the capacitor entirely of materials which Jun e Aery small mechanical temperature coefficients. Then provided that the electrode distortions and dis- placements caused by temperature dependences of

t lie elastic coefficients of the electrodes and electrode

supports are not significant, the capacitance temperature coefficient will also be small. Materials of small

mechanical temperature coefficient which have been

used for the construction of capacitors are fused

silica and Invar. Fused silica is difficult to form

and mount, and must be metalized to provide elec- trodes. Its exclusive use in a capacitor with air dielectric results in temperature coefficients of capacitance around 0.5 ppm/°C. Invar is somewhat

unstable and must be carefully heat treated. When

treated for optimum stability, temperature coeffi- cients around +2 ppm/°C are typical.

The disadvantages of fused silica and Invar are often avoided by the use of two or more materials of

different but relatively high temperature coefficients

in an arrangement that provides temperature compensation. Temperature compensated capacitors

using two metals have been built in many ways and

with varying degrees of success. One simple but

somewhat unsatisfactory design makes use of coaxial

cylinders in which the outer cylinder has a higher

temperature coefficient than the inner. The dimensions of the capacitor are chosen so that the capacitance between the cylinders is independent of temperature, provided that no temperature differential

exists between the two cylinders.

Since the temperatures of the various elements in

a capacitor are not generally equal but vary with

the room or bath temperature with possibly different

time lags, quite large temperature differentials and

capacitance changes can result. The cylindrical

design is particularly objectionable from this standpoint, since the outer cylinder is often in more

intimate contact with the surroundings than the

inner cylinder, and hence responds to temperature

changes more rapidly.

A temperature compensation technique often used

for capacitors larger than 100 pF involves stacking

plates between spacers having a higher temperature

coefficient than the plates. A second similar stack

of plates is interleaved with, but insulated from, the

first set of plates. Since the dissimilar metals are in

intimate thermal contact, temperature differentials

between plates and spacers remain relatively small,

and the capacitance between the two stacks is not

strongly affected by tempeiature fluctuations.

A more elaborate system of temperature compensation may be developed by considering the effect on

the direct capacitance caused by independent

variations in the temperatures of the separate

electrodes. This procedure is found to be most

useful for capacitors smaller than a few picofarads

in which the temperature of the ground electrode

may also be important.

2. Partial Capacitance-Temperature

Coefficients

Consider a three terminal capacitor whose direct

capacitance is given by C=C(tlt t2 , t3 ) where the t's are the temperatures of the three electrodes. Then

for small departures from the reference temperature,

5C dC

'<Ma

i »r C=C'o(l +

\_ dC J_ dC

Co d*2 J_ dC

C0 dt 3 = C0 (l+a1 A/ 1 +a2Af 2 +a3Ay (1)

504
The terms ^~=ai etc, are the partial capacitancetemperature coefficients of the three electrodes.

If we let /, be a characteristic length of electrode 1, we may write

«1 dfx We recognize 2 = /3 as

the thermal expansion coefficient of electrode 1, and write

a!=Pi/3, where Pi=^r tr5 etc., and obtain

C

7T=l+P1 /3 1 A* 1 +P2 /32 AM-P3&Af3

. (2)

If /3, =jS2 =/33 and Atl = M2 =At3 , we know from

geometrical considerations that

C

1 +iSAf, and

hence I\ + P2 + P3 = 1, but no other restrictions on

the P's exist. Equation (2) suggests the possibility of designing

capacitors with P2 =Pi =0 so that P, =l. Tlien if

j3, is small, C will be relatively independent of Atu At2 , and At3 . This can be done by constructing

electrode 1 of fused silica and choosing a suitable

geometry. Such a capacitor may be adjusted by

machining operations on electrodes 2 and 3, which

may be of any stable metal, and the principle objection to the use of fused silica is eliminated.

The above analysis is correct if no temperature

gradients exist in any electrode, and if the capaci- tance changes caused by motions of the electrodes

with respect to each other are not appreciable. The

first criterion can be partially satisfied by avoiding

the use of long, thin electrodes. The second cri- terion can be satisfied by locating the electrodes at saddle points in the capacitance-displacement function. Experimentally it is often convenient to set one

of the P's, say P2 =0, by a geometrical consideration,

and then either to adjust P3 =0 or Pj = l by observing the capacitance changes brought about by

changing the dimensions of electrode 1 or electrode :l. The easiest approach will be determined by the

capacitor configuration

.

3. Guard-Cylinder Capacitor

Consider a guarded cylindrical capacitor whose

inner electrode has a radius a, and whose outer

electrode and guards have a radius a2

. If the length

of the outer electrode is / and if e is the permittivity

of the dielectric^ the direct capacitance of this system

is given bv

2irel

In (I)

Differentiation yields

c = 1

1 Aa, 1 Aa2 , Al

(4)

For a uniform expansion of the outer cylinder,

Ao2 Al a2 I and C is independent of the temperature of

the outer cylinder if In — =l.or—=e. This relationship results in temperature compensation for the

outer cylinder, but only if the guard expands at the same rate as the guarded electrode; and results in a capacitor whose temperature coefficient of capaci- tance is equal to the thermal expansion coefficient of the inner cylinder.

Although independent compensation for guard and

guarded electrode temperature changes cannot be

achieved with the simple guard cylinder capacitor, some insight into the behavior of more complicated

cylindrical systems is derived from the results obtained. The critical ratio of radii, which is e for the

simple guard cylinder capacitor, is found to be

quite close to this value for a large variety of completely compensated cylindrical designs.

4. Practical Compensated Capacitors

After some model studies, an experimental 5 pF

capacitor was constructed as shown in figure 1. The inner electrode was a metallized fused silica cylinder. The guard electrodes were set back as

illustrated to reduce the influence of their temperatures upon the direct capacitance of interest. It was

found experimentally that small changes in the

length of electrode 2 did not appreciably affect the

compensation, so that adjustment to nominal value

is simple.

Tests of the completed capacitor showed an overall

temperature coefficient less than 1 ppm/°C and no

appreciable overshoot when subjected to a rapid

temperature change.

wzzzzzzzzzzzzzm h Y/77/////////////77. ^ELECTRODE I L

ELECTRODE 2 ^

FKM'RE 1. An experimental 5 pF completely temperature

compensated capacitor drawn roughly to scab for correct compensation, a-ila\ ~2.5.

505
It is found that the design of figure 1 is not easily

modified to yield either much larger or much smaller

capacitance. Other designs have been investigated,

however, which may be useful for the construction

of arbitrarily small completely compensated capacitors. One such design is sketched in figure 2. It makes use of a cylindrical shield between the two

capacitor electrodes. The inner electrode is once

again constructed of metalized fused silica. Holes in the cylindrical shield may be varied in sizes to provide a range of capacitances between the inner and

outer electrodes. The outer electrode is automatis-^ELECTRODE I- <Z ELECTRODE 2 // // / / /

°3

o>

r

FlGURE 2. A completely temperature compensated capacitor for

values smaller than 0.1 pF.

cally temperature compensated by virtue of its shape.

Any increase in its temperature will increase the

spacing between it and the cylindrical shield, but will not change the position of the critical conical surfaces. The outer ends of the cones must of course be truncated, but it is found that this can be done

without greatly upsetting the compensation. Experimentally it is found that the critical ratio of cylindrical shield radius to inner electrode radius is somewhat dependent upon the size and shape of the

holes in the shield, but remains remarkably close to e

\

5. Conclusions

Most practical capacitors are found to be not

greatly dependent upon temperature differentials. However, complete disregard for the principles of partial temperature compensation can result in capacitors which are unduly sensitive to rapid temperature changes. It is possible at least in principle

to design capacitors which depend only upon the

temperature of one element ; and if this element has

a small thermal expansion coefficient, very close temperature control is not necessary. The technique seems most practical for capacitors smaller than

10 pF.

(Paper 6SD4-177)

506
NATIONAL BUREAU OF STANDARDS

technical ^ote 201

Issued November 5. 1963

A TECHNIQUE FOR EXTRAPOLATING THE 1 KC

VALUES OF SECONDARY CAPACITANCE STANDARDS

TO HIGHER FREQUENCIES

R. N. Jones

Radio Standards Laboratory

National Bureau of Standards

Boulder, Colorado

NBS Technical Notes are designed to supplement the Bureau's regular publications program. They provide a means for making available scientific data that are of transient or limited interest. Technical Notes may be

listed or referred to in the open literature.

For sale by the Superintendent of Documents, U. S. Government Printing Office Washington, D.C. 20402

507
Contents

Page

Abstract 1

1. Introduction .1

2. Procedure for Determining High Frequency Value 3

3. Sources of Error 7

4. Conclusion J.1

5. References 12

508
A TECHNIQUE FOR EXTRAPOLATING THE 1 KC VALUES

OF SECONDARY CAPACITANCE STANDARDS TO

HIGHER FREQUENCIES

A simple technique is described for extrapolating the

1 kc values of certain two -terminal capacitors to higher

frequencies without incurring serious losses in accuracy.

The method is intended for use with air capacitors having

binding post or banana plug type connectors. Because of

inherent errors, such connectors are not appropriate for

measurement where the highest accuracy is required. For

this reason, it is recommended that calibration of this

type be performed by secondary laboratories and not be

submitted to the National Bureau of Standards

.

1. INTRODUCTION

With the establishment of requirements that military contractors

show traceability to the National Bureau of Standards in all areas of

measurement, increasing demands for new and improved types of

calibration services have appeared. Included in these demands is

the entire area of high frequency impedance measurements. Prior

to the increased demand for traceability there was little requirement

for impedance standards and calibration services at frequencies much

above 1 kc, and as a result there existed almost no commercially

available standards which were designed for high frequency applications. Consequently, NBS has been requested to perform high

frequency calibrations of standards which were primarily intended

for use at dc and low frequencies. Such requests have presented

difficulties because NBS does not possess appropriate standards to

use for comparison, nor is it practical to attempt to develop them.

A limiting factor in the accuracy of impedance measurements is

associated with the connector systems which are employed as a part

509
of the instrumentation set-up. This is true to a varying degree at

all frequencies, but in general, as the frequency of measurement is

increased, the importance of connectors must be given added consideration. There are certain minimum requirements which should

be satisfied by any connector system in order to assure accurate

results in impedance measurements. First, the connector should

be coaxial for purposes of self -shielding so as to avoid any influence

of surrounding objects upon the measurement result. This requirement is an important one even at the very low frequencies, especially

when small values of capacitance are to be determined. Two other

requirements are a well defined plane of electrical reference and a

voltage standing wave ratio (VSWR) as nearly 1 . 0 as possible. These

are most important at the higher frequencies where short wavelengths

are encountered and measurement accuracy depends upon the degree

of impedance matching within the connector.

Many of the impedance standards submitted to NBS for calibration

at frequencies above 30 kc employ banana plug connectors and are,

therefore, subject to connection errors which render them inappropriate

for use where highest accuracy is desired. Of particular interest

here are capacitance standards which are equipped with banana plug

connectors of either the plug or jack type.

Because the calibration methods employed at NBS for standards

of this type do not depend upon any national reference standard, it is

possible for a laboratory to perform its own high frequency calibrations

of such capacitors and avoid the necessity of submitting them to NBS.

By employing accepted techniques to calibrate such secondary standards, mutual benefits can accrue both to NBS and to the laboratories

requiring calibration services. NBS, bv limiting services to only the

510
highest quality standards, can devote increased effort to research

and, as a result, provide better services. On the other hand, other

standards laboratories can save money and time and avoid the perils

of transporting standards by utilizing as many such self-calibration

techniques as possible.

The measurement technique to be described is not new, nor is it

complicated. It is presented as an aid to overall efficiency, and while

it does contain several sources of error, they are not out of reason

when compared with the general accuracy to which standards to this

type may be considered reliable.

2. PROCEDURE FOR DETERMINING

HIGH FREQUENCY VALUE

Let us assume that we have a two -terminal air capacitor, with

binding post or banana plug connectors, whose known low frequency

(1 kc) value in farads is CQ, and that we have an application requiring

that we know the capacitance value, Ce , at some higher angular frequency, go = 2-rrf, where f is in cycles per second. Inherent in any

capacitor is a certain amount of conductance, series resistance, and

series inductance. All of these contribute to the effective value of

the capacitor at any given frequency, but, in this case, it is only the

series inductance which is considered significant. This is especially

true for high Q capacitors. Consider the simplified equivalent circuit

for the capacitor as shown on the right in figure 1.

Ce Co

Figure 1 Simplified Equivalent Circuit

511
Here L. is the residual series inductance of the capacitor in

henries and CQ

is the low frequency (1 kc) value of the capacitor in

farads. At some angular frequency, we wish to find the effective

value Ce . Equating the reactances of these two circuits gives:

1 1 T

and

c

Ce = § (1)

l-oo JLC

Q

To arrive at a value for C

g

, it is necessary to evaluate the residual

series inductance, L. This is easily done by short circuiting the terminals of the capacitor and employing a grid -dip meter to ascertain the

angular frequency, oj^ = 2-rrfr , at which the resulting circuit resonates.

The residual series inductance of the capacitor is then given by the

expression:

L=-| Ls (2) r o where CQ

is again the low frequency value of the capacitor and L. g is the inductance of the device used to short circuit the terminals.

The effective capacitance is then obtained by substituting the value

for L into (1). A typical set-up for determinations of this type is

shown in figure 2.

The solid line in figure 3 indicates the percentage correction, as a function of frequency, for a 1000 picofarad capacitor having a residual series inductance of 0.050 microhenry. This is a typical

value for the inductance of a good capacitance standard and may be

expected to be nearly the same for all capacitors of the same general

type of construction and comparable physical size. With this in mind,

512
Figure 2

Grid -Dip Meter Arrangement Used to Determine the

Resonant Frequency of a Short-Circuited Capacitor.

513
100

100 tc/s I Mc/s 10 Mc/s PERCENT CORRECTION VERSUS FREQUENCY FOR A IOOO pf CAPACITOR

HAVING A RESIDUAL INDUCTANCE OF 0.050 MICROHENRY

Figure 3

514
it should be noted that the larger the value of the capacitor being

considered, the more the effective value will rise for a given frequency.

3. SOURCES OF ERROR

Four principal sources of error associated with this measurement

procedure are as follows: (a) inaccuracy in the determination of the

resonant frequency of the short-circuited capacitor, (b) mutual

coupling between the shorting device and the grid-dip meter coil,

(c) the inductance of the shorting device, and (d) the equivalent circuit used to represent the capacitor. In examining these sources of

error it is necessary to generalize in a number of instances because

of the absence of a convenient geometrical arrangement upon which to

base a rigorous analysis. This is especially true in the latter three

instances where current distributions in conductors and the resulting

fields are extremely complex and true representations of actual circuit conditions are almost impossible to realize.

In measuring the resonant frequency of the short circuited capacitor, the grid-dip meter indication is very distinct because the capacitor

and short constitute a high Q circuit. This condition makes in possible

to achieve repeatability which is beyond the resolution of the frequency

scale on the grid -dip meter oscillator. Either of two methods may be

employed to improve this resolution. The frequency of the radiated

signal of the oscillator may be measured with a hetrodyne frequency

meter, or the signal may be amplified and fed into a frequency

counter. In this manner, it is possible to achieve repeatability well

within one percent. Assuming that the accuracy is also of this order,

the effect on the accuracy of the total circuit inductance would be

approximately two percent. For frequency accuracies within a few

515
percent, the accuracy in inductance is given by the expression:

—AL Ago — re

-Li U)

where AL, and Aco represent the errors in inductance and angular frequency respectively.

The error due to mutual coupling between the short and the griddip meter coil is evidently very small because no distinct change in

the resonant frequency is caused by varying the distance between them.

Any change in the resonant frequency resulting from an increase or

decrease in the separation is probably of the same order as the stability

and setability of the grid-dip meter oscillator. In any event, the

coupling error may be minimized by maintaining a separation of an

inch or more between them.

Probably the largest source of error is in the value used to

represent the self -inductance of the short. It is possible to improve

the situation by making' a judicious choice of the size and configuration

of the device used so as to permit the use of available formulas

[ Terman,1943] to calculate the self -inductance . Even so,there

remains a margin of uncertainty due to the geometry of the current

path, and such formulas may not be expected to yield accuracies

better than ten to fifty percent. However, since the inductance of

the short is usually only a small portion of the total inductance of

the resonant circuit, this does not represent a serious degradation

of the value obtained for L in equation (2). Figure 4 is a photograph

showing two shorting bars which are convenient to use. These are

brass bars fitted with banana plugs or jacks appropriately spaced

to fit the capacitor terminals.

516
517
Finally, we must consider the equivalent circuit used to represent

the capacitor. The circuit of figure 1 may be criticized because, in

actuality, the inductance and capacitance probably do not exist as shown. Depending upon the physical details of the construction, the

capacitor may be more accurately represented by other equivalent

circuits. As an example, consider the equivalent circuit of figure 5.

Figure 5

High Frequency Equivalent Circuit

Here the inductance and capacitance, instead of being lumped together

at one point, are separated into fractional parts of the total and

arranged in a manner which more nearly approximates the actual

situation. Ct is the capacitance between the case and the high

potential lead passing through it, and Cp is the capacitance in the

stack of plates. L^. and are the lead inductances arranged as they

may appear in relation to C^. and Cp. Problems arise, however,

because it is difficult to estimate the values to assign these components,

and also because complex circuits are cumbersome to handle mathematically. To illustrate the importance of the equivalent circuit,

consider a capacitor having a 1 kc value of 1000 picofarads and a residual series inductance of 0.050 microhenry. Using the equivalent

circuit of figure 1, the value at 1 Mc is 1001.98 picofarads. The

circuit of figure 5 gives a value of 1001.96 picofarads if we let

Lj. = Lp = 0.025 microhenry, Ct = 10 picofarads, and Cp = 990 picofarads. At this frequency the difference is essentially negligible,

518
but at 15 Mc the values are 1799.00 picofarads for the circuit of

figure 1 and 1786.32 picofarads for the circuit of figure 5, using

the same values for Lu., L , C., and C . Thus it is seen that the t* p' t p

equivalent circuit assumes increasing importance at the higher

frequencies

.

Because of the sources of error, it is believed that the reliability

of the value obtained for the residual series inductance by the method

described is not better than ten percent. Again referring to figure 3,

the broken lines indicate the upper and lower limits of the corrections

which result from this ten percent uncertainty in the inductance value.

In general, it may be said that the effective capacitance may be determined by this method to accuracies of one percent or better up to

frequencies where the correction amounts to ten percent of the 1 kc

value

.

4. CONCLUSION

With the advent of improved rf connector systems, many standards

and instruments using the older type connectors are no longer appropriate for use at the level of primary standards. NBS has developed and

does maintain standards which are only appropriate for use with the

improved type of coaxial connectors [Jones and Nelson, 1962] . Because

it is impractical for NBS to attempt to maintain standards for a wide

variety of connector configurations, secondary standards laboratories

are urged to employ this calibration procedure where possible and

when compatible with traceability requirements.

519
5. REFERENCES

1. Jones, R. N., and R. E. Nelson (1962). "The Role of

Capacitance in the National Reference Standards of

High Frequency Impedance, " Proceedings of the

Instrument Society of America, Paper No. 18.1.62.

2. Terman, F. E. (1943). Radio Engineers Handbook, First

Ed., pp. 48-51 (McGraw-Hill Book Co., Inc., New York,

N. Y.).

520

GPO 837- 107
Four-Terminal-Pair Networks as Precision

Admittance and Impedance Standards

R. D. CUTKOSKY

MEMBER IEEE THE DEVELOPMENT of suitable standards for the maintenance of impedance and admittance has been along two separate lines. One of these lines is based on the concept of 4-terminal measurement, which requires a standard

to be provided with two current and two

potential terminals. This technique has

long been used with low d-c resistances,

and may be applied directly to the measurement of small mutual inductances. The technique results in the elimination of uncertainties caused by variations in contact impedances. The other line of ap- proach is applicable to very high impedances in which case even a very low stray

admittance between the terminals could change appreciably the apparent impedance of the standard. The effect of stray

admittances is eliminated by shielding the standard and making measurements in such a way that stray admittances to the

shield are not included in the measurement. The technique is most often applied to the measurement of low capacitances; this standard is called a 3-ter- minal capacitor. The technique is also

feasible for use with very high d-c resist- ances. An interesting point is that if neither 4-terminal construction nor shielding is

Paper 63-928, recommended by the IEEE Fundamental Electrical Standards Committee and

approved by the IEEE Technical Operations Committee for presentation at the IEEE Summer

General Meeting and Nuclear Radiation Effects Conference, Toronto, Ont.. Canada, June 16-21,

1963. Manuscript submitted March 18, 1963; made available for printing April 26, 1963. R. D. Cutkosky is with the United States Depart- ment of Commerce, National Bureau of Standards, Washington, D. C.

provided, the most reliable measurements

are made in a middle range of impedance

in which errors due to contact impedance and shunt admittance are about

equal. However, precision attainable in this range may not be as high as desired. The provision of current and potential terminals results in an improvement of precision for very low impedances, with very little improvement in the middle- and high-impedance range. The use of shielded or 3-terminal construction results

in an improvement of precision for very

high impedances, with very little improvement in the middle- and low-impedance

range. Neither system alone substantially improves the measurement precision in the middle-impedance range. The purpose of this paper is to develop a theoretical basis for measure- ment techniques involving both shielding

and current and potential terminals, and

to show that the generalized impedance

standard so defined reduces to the 4- terminal impedance when it is very low

and to a shielded (3-terminal) admittance

when its impedance is very high. Partial solutions to the problem have been de- scribed in the literature, 12 but without

sufficient attention to the details of ground connections. Some concepts and

techniques of modern network theory are used in the analysis. 3

Refinement of 3-Terminal Capacitors

For many years 3-terminal capacitors

have been widely used as low-value capaci- tance standards. Three-terminal construction greatly reduces errors due to stray and variable capacitances between

the terminals. Careful analysis of the behavior of such devices shows that, al- though the uncertainty caused by variations in lead geometry is rather small, it is not negligible when the capacitances are

high, or if extremely high accuracy is desired. Three-terminal admittances are usually constructed with two coaxial connectors

as terminals, and with the outer members

of the coaxial connectors joined with a low-impedance conductor. Since this conductor has a finite impedance, the standard has in reality four terminals,

and its current response to applied volt- ages must be described by the six in- dependent elements of its three-by-three

admittance matrix. To reduce the number of variables, it is common practice to

treat the standard as a 2-terminal-pair device, whose current response to voltages

at the pairs of terminals may be described with only three independent admittances.

This approach assumes first, that we are

interested only in the loop currents at the 2-terminal pairs caused by voltage applied across these pairs; and second, that

either currents in all other external loops

are identically zero or, equivalently, that

Fig. 1 . Two-terminal-pair admittance

Fig. 2. Two-terminal-pair admittance with

series lead impedances

521
(A)

z,

i—r^-h—

: [ft [ft

:

z,

7] z2 :

PAIR

3,!

PAIR 2( >PAIR4,

(B)

Fig. 3. Four-terminal-pair admittance A—As constructed B—Simplified circuit

the current into the inner member of each coaxial connector is exactly equal to the current out of its outer member.

Mathematically, this approach allows the impedance between the coaxial connector's outer members to be suppressed when

the device is represented by an equivalent

circuit. An equivalent circuit for a 2-terminalpair admittance is shown in Fig. 1. If

and

t2 = «l K2 l+e2 K22

then Y»=YA +YT Y22 = Yb + Yt

and Yi2 = Y2l =-YT

Fig. 1 is clearly similar to the usual equivalent circuit for a 3-terminal ca- pacitor, in which YT =juC, where Cis the

direct capacitance. This analysis implies that there may be

conditions under which an "ordinary" measurement of a 3-terminal capacitor

may be uncertain by an amount dependent upon net current from one coaxial terminal pair to the other, and upon the

internal construction of the standard.

It is thus suggested that the transfer ad- mittance of a 2-pair network

V -*\ y21 —

6l!«2 = o may be a more precisely defined parameter

than the loosely defined 3-terminal admittance.

If net current between terminal pairs

is suppressed and if all leads are coaxial, the current then also will be coaxial and

no magnetic fields will exist outside of the cables. This astatic condition reduces

the errors that otherwise might be pro- duced by stray fields. Carried to an

extreme, one can see that the admittance

standards themselves should be as astatic as possible. Although this condition is usually fairly difficult to achieve, it may be improved by the use of eddy currents or magnetic shields, as they are needed. When lead impedances becomes signifi- cant, the network of Fig. 1 must be modified as in Fig. 2. An analvsis shows that

ifYu= YA+YT Ym = YB+YT and

F,2= Y,x = - Yr =y2i'=

l+Zl Yll +Z2 Yi2 +Zl Z2 ( Yu Y22 - F12 2

) (i)

If all terms of the form (Z 1') are much

smaller than unity,

If Zt = Z2 = 10

2 ohms and Yn = FM =1()-4 mhos, the expression Z\Y\\-\-Z2 Yn repre- sents an error term in the measurement

of Y2\ with a value of two parts per

million. In practice, it is found that one may

work conveniently with Yn

' instead of Y21 for intercomparison purposes. Even

if Yn and Y& are constant, accuracy is limited by the repeatability of Zi and Z2 . These impedances include, for the system under study, the resistance of the coaxial connectors used to make contact with the standard; for good connectors, they will be uncertain by less than 10~4 ohms. If Yu and F~22 are about 10 -4 mhos, Y2l

' will be measurable to a few parts in 10s

. Special connectors could no doubt be

constructed to reduce the contact uncertainty, but if Yn is very high, a limit

will be reached beyond which uncertainties due to lead variations cannot be re- duced conveniently. This difficulty may be circumvented by

the use of 4-terminal techniques, such as those used for low-value standard resistors. In our case, the 2-pair network

is then replaced by a 4-pair one. Some

properties of general 4-pair networks are derived in the Appendix, but these properties unfortunately do not have a simple

physical interpretation. One logical extension of the 2-pair net- work can be constructed by extending and

splitting each coaxial pair with a per- manently attached coaxial T-connector,

Z 3

[ft ft ft ft

2

Fig. 4. Four-terminal-pair admittance with two pairs ignored

as in Fig. 3(A). Using suitable Ydelta transforms, this may be simplified to

Fig. 3(B).

If we constrain i2 = 0, u = 0, and et = 0

in Fig. 3(B) by placing suitable generators

at terminal pairs 1 and 3, we find the

following equation results:

= Yt {l+Y3 Zt )(\+Y2Z2 ) (2)

which we define to be the 4-pair ad- mittance P3214

. Z2 and Z3 will typically be the impedances of a few inches of coaxial line, and Y2 and Y3 the shunt admittances of the same lines. Hence F2Z2 and Y3Z3 will be very small at audio frequencies. Numerically, we may

expect that if Z2 = Z3 = 10~2 ohms at Y2 =

73 =10-7 mhos (10 picofarads if w=104 radians/second), then Y2Z2 =Y3Z3 =\§~'3

. If Z2 , Z3, Y2 , and Y3 are each constant and repeatable to 1%, P32 14 will be defined with a connection uncertainty of a few parts in 10 n. Lastly, it should be noted that a very

small 4-pair admittance for which all (YZ) products are negligible reduces to the normal 2-pair transfer admittance if terminals 2 and 4 are ignored (open cir- cuit). In this case Fig. 3(B) may be redrawn as Fig. 4. The transfer ad- mittances of this 2-pair circuit may be

calculated to be

1+Z,( YT+ Yx )+Z3 ( Yt + Yr )+

Z,Z3 ( YT Yx +YT YY +Yx Yr )

(3)

Fig. 5. Two-terminal-pair impedance

Fig. 6. Two-terminal-pair impedance with shunt lead admittances

522
O ! PAIR 1 o ^~

z, Zj_[§] PAIR 2

t

1 o VPAIR 3 o J—

z3

A/\—--j O

24

[jfl] PA1R 4

Fig. 5 represents a general 2-terminalpair (4-terminal) impedance. We may

write

Fig. 7. Four-terminal-pair impedance

Fig. 8. Special 4-terminal-pair impedance

constructed with a mutual inductor

where

and

YY = YB+

l+Z,Yt

In the limit as all YZ products become

negligible, equations 2 and 3 reduce to the following:

P*u=-Yn =Yr

Refinement of Mutual Inductors

Four-pair networks may be constructed

in many ways for use at low-impedance

levels. A complete survey of possible construction practices will not be at- temped here. Discussion will be limited

to one special case in which the 4-pair

impedance has the highly desirable property of reducing in the low-impedance

limit to the ordinary 2-pair transfer impedance.

where Zl2 = Zn = ZT

is called the transfer impedance. As with the 2-terminal-pair transfer admittance of Fig. 1, net current from one

pair to the other of the network must be

suppressed or its effect on the apparent

transfer impedance investigated. Otherwise, Fig. 5 is not a valid equivalent cir- cuit.

If the shunt admittances of the leads

are taken into account, Fig. 5 must be

replaced by Fig. 6. Analysis shows that

*2' = 0=Zn'

1+ YaZ»+ YBZn+ YA YB(ZuZn - Zn*)

(4)

If all YZ products are small, so that terms of the form [YZ) (YZ) are negligible, we have Za'—

2

a (l- YAZn - YBZa). We may work with this Z2/ instead of Zi\ for intercomparison purposes, but ac- curacy is still limited by the repeatability

of YA, YB, Zn , and Z22. The arguments

are similar to those associated with

equation 1 , which resembles in form equation 4. Elimination of errors due to shunt ad- mittance may be achieved by the use of shields around each individual terminal. This may be compared with shielding a

2-terminal capacitor to produce a 3- terminal one. Fig. 7, which shows such

a shielded transfer impedance, is identical with Fig. 3(B) if

1 Y^zT We may hope to use the same conditions

£2 = 14 = 0 and e4 = 0, so that

iV' (1+Z1 Y2 ){1+Z3 Y3 ) (5)

Fig. 9. General 4-terminal-pair network

and find a meaningful interpretation for Qa1 * in the limit as all Z including ZT be- come small. That this may be done for the special case of a mutual inductor will now be shown.

Consider a mutual inductor constructed

as in Fig. 8. For this circuit

(?3!

l4 = d+z2 y2 )(i+z3 F3 ) (6)

If terminal pairs 3 and 4 are short-cir- cuited, we may calculate the mutual im- pedance between the remaining terminal

pairs 1 and 2 to be

= Z2I «i i2 = 0

{\+Z1 Yi)[\+Yi{Zl +Zx)]-Ztl Yl Y1(7) where

Zj- z3 z, Z3 +Z,+Z3 Z,YA

If all YZ products are negligible, equations 6 and 7 reduce to Z21 = Qzi lt = ZT , the

desired correspondence.

General Considerations

It has been shown that it is theoretically

possible to construct 4-terminal-pair devices having properties dimensionally

equivalent to admittance or impedance,

and that these suitably defined properties are relatively independent of the properties of the leads connected to the net- work. The technique is expected to be

useful as an aid in designing a wide range

of impedance standards which may be intercompared and used to calibrate other

standards. Four-pair standards will have

the same physical meaning e,t very low and very high impedances as presently

existing standards. In the middle-im- pedance range, they may be used as well- defined elements in an impedance step- up chain. They may also be used to measure ordinary 2-, 3-, or 4-terminal standards with accuracies limited only by

the ordinary standards. Construction of 4-terminal-pair de- vices may follow the existing techniques

for constructing ordinary standards.

Capacitors, resistors, and self- or mutual

inductors are expected to be useful. Regardless of the construction, 4-pair adO -j— —j—

ij Me,| i,) —p^W—

|— 0

z 2

f. 52 z«

2] ^(^P^e*! £)

LU lJ- 2 U-0 ([4 \e*\u\ [y[]@, |<

Fig. 10. General 4-terminal-pair network

with lead*

523
mittances must be carefully designed to be astatic; that is, they must neither produce stray field nor be susceptible to pickup from existing stray fields. This is particularly important with very low

impedances. Search coils may be used

to investigate standards for this effect. The preceding discussion has centered around a particular type of 4-pair net- work. Unfortunately, the conclusions do

not hold for general 4-pair networks. For this reason, it is essential to test a completed standard to determine whether

or not the required conditions of this paper can be met. Varying lead lengths and

noting the result may be one test The

appendix, which treats a general 4-pair

network, may be used as an aid in de- veloping appropriate tests. Just as im- portant is an experimental determination

of the effectiveness of the methods used

to suppress unwanted loop currents be- tween terminal pairs.

It is not the purpose of this paper to describe techniques for measuring 4- terminal-pair admittances. However, ex- perimental work has been done to verify some of the conclusions made here. Measurement techniques will be discussed

in a subsequent paper.

Appendix

If the network of Fig. 9 is linear, we may write

h = «i Fn + e2 Fi2 4-e3 Fi3+e4 Yh

k = e, Y2l +

2 F22 +3 F23 +4 F24

k = ei F31+2 F3 2+e3 F33 +e4 F34

ii = el Y„ +

2 Y,2 +

3 F43 +e,Y„

If we require that e4 = 0, k = 0, and

i 4 = 0, we find — k

Yn +

YvYuYu+YnYuYnY,l Yi2 Y2l -Yzl Y12 Yil

Yv. Yw — F43 F2 i

which we define to be the 4-pair admittance

Pit 1 *. It is seen that if reciprocity holds, iVl = iV4

. An expression for the error due to series impedance and parallel admittance of leads may be calculated by considering the circuit of Fig. 10. A calculation yields for k' —

i4

' = 0 and e/ = 0.

^=(/v«)' =

iV4 +

where

(F31 F43 - Ym Ya )YA +{Y2l Yi2

- YuYvWb-YuYaYb

F43 F21 — F41 F23 K

Yi+Yt'+YiYSZt

I+Z2F2'

_

F3 +F3 '+F3 F3 %

" \+Z3 Y2

'

and K = (l+Z2 Y2 ')(1+Z3 Y,')

This is of a form which may be treated experimentally by varying Ya and Yb and

observing the change which is brought about

in (/V4 )'.

References

1. A Dbsign for Resistors of Calculable AC/DC Resistance Ratio, D. L. H. Gibbings.

Proceedings, Institution of Electrical Engineers, London, England, vol. 110, Feb. 1963 pp. 335-47.

2. Evaluation of the NBS Unit of Resistance Based on a Computable Capacitor, R. D.

Cutkosky. Journal of Research, National Bureau

of Standards, Boulder, Colo., vol. 65A, no. 3, May-June 1961, pp. 147-58.

3. Introductory Circuit Theory (book), E. A. Guillemin. John Wiley and Sons, Inc , New York,

N. V , 1953.

A reprint from COMMUNICATION AND ELECTRONICS, published by The Institute of Electrical and Electronics Engineers, Inc. Copyright 1964, and reprinted by permission of the copyright owner The Institute assumes no responsibility for statements and opinions made by

contributors. Printed in the United States of America

January 1964 issue

524
621.319.42 The Institution of Electrical Engineers

Monograph No. 35 1 M

Jan. 1960 ©

SOME RESULTS ON THE CROSS-CAPACITANCES PER UNIT LENGTH OF

CYLINDRICAL THREE-TERMINAL CAPACITORS WITH THIN DIELECTRIC FILMS

ON THEIR ELECTRODES

By D. G. LAMPARD, M.Sc, Ph.D., and R. D. CUTKOSKY, B.Sc.

(The paper was first received \st April, in revised form \llh July, and in final form \%th August. 1959. // was published as an Institution Monograph in January, 1960.) SUMMARY

The effect on the cross-capacitances per unit length of cylindrical 3-terminal capacitors of thin dielectric films on the capacitor electrodes

is discussed.

It is assumed that the cross-section of such dielectric films remains

constant throughout the length of the capacitor. Some conformal

transformations of basic cylindrical capacitor cross-sections are given. When these are applied to a symmetric cylindrical capacitor with a

thin dielectric film on its electrodes, the results suggest that the mean

capacitance per unit length may remain constant to the first order

despite the presence of the dielectric film. The same methods also suggest that the individual cross-capacitances per unit length may

remain constant to the first order provided that the dielectric film is disposed symmetrically with respect to the capacitor symmetry plane. Further support for these conjectures is given by the results of a

detailed calculation of the cross-capacitances per unit length of a

parallel-plate cylindrical capacitor with a thin uniform dielectric film on one electrode. In the last Section, a lemma concerning the existence of an equivalent dielectricless capacitor is given and this is followed by

the proof of general results of the type suggested by the previous

working.

(1) INTRODUCTION

A new approach to the design of calculable 3-terminal capaci- tance standards has recently been made possible as a consequence

of a general theorem1 concerning cross-capacitances of certain

cylindrical electrode systems. This theorem enables a capacitor

to be constructed such that only one precision length measure- ment is necessary to compute its capacitance to a high order of

accuracy. A proof of this general theorem, together with detailed calculations of cross-capacitances of cylindrical systems with a

variety of cross-sections, has been given in a recent paper. 2 In

that paper the effects of various perturbations which could occur

in practice were considered, but an important practical perturbation, namely the presence of spurious thin dielectric films on the capacitor electrodes, was dealt with only in one special case. Such dielectric films can be present, for example, in a precision calculable cylindrical capacitor, where it may be difficult to avoid

surface contamination of the electrodes with oil films, adsorbed

vapours, etc. They can also arise in connection with the direct experimental determination (by capacitance measurement of a

cylindrical capacitor) of the permittivity of vapours, where a

thin film of condensed vapour on the capacitor electrodes may

constitute a serious source of error. In the paper we examine further the effects of thin dielectric films in cylindrical capacitors, the assumption being made that

the cross-section of the dielectric film remains constant throughout the length of the capacitor. To facilitate the discussion of

Correspondence on Monographs is invited for consideration with a view to publication. Dr. Lampard is in the Division of Eleclrotechnology of the Commonwealth Scientific and Industrial Research Organization. Australia, and Mr. Culkosky is at the National Bureau of Standards, Washington, D.C., United States.

these dielectric effects, a number of conformal transformations

are given in Section 2 by means of which a cylindrical capacitor of arbitrary cross-section can be mapped into a cylindrical

parallel-plate capacitor. This procedure provides a simple

alternative proof of the main theorem on cylindrical capacitors

in a more generalized form.

In Section 3 it is shown how some uniform dielectric distribu- tions behave under the transformations of Section 2, while in Section 4 both cross-capacitances per unit length of the cylindrical parallel-plate capacitor of Section 2, with a highly asymmetric

dielectric distribution on its electrodes, are calculated in detail. The calculations of Section 3 and 4 suggest some general results, which are established in Section 5. concerning the effect of thin

dielectric films.

(2) GENERALIZATION OF THE THEOREM ON

CYLINDRICAL CAPACITORS

Consider a conducting cylindrical shell whose right crosssection, Fig. 1(a), is the arbitrary closed curve S. Let this shell be divided into four parts by infinitesimal gaps at a, /?, y and 8. By means of an existence theorem due to Riemann,3 in the theory of conformal transformations, the interior of the cross- section 5 can be mapped into the interior of a circle, Fig. \(b), and with a suitable choice of a co-ordinate system the point a

can be mapped on to the origin and the centre of the circle taken on the real axis. The mapping to = 1/z transforms the interior of the circle of

Fig. 1(b) onto the right-hand half-plane of Fig. 1(c), where the distances |9y and y& have been arbitrarily called b and a, respectively. By means of simple rotations and translations, the right half- plane of Fig. 1(c) can be mapped into the upper half-plane of

either Fig. lWXO or (ii). Finally, the mapping v = logs u transforms the upper half- planes of Fig. 1(d) into the strip spaces of Fig. 1(e). The configuration of Fig. \(e)(i) is that of the usual guarded

parallel-plate capacitor. The capacitance per unit length can be computed by assuming electrode a/J to be at unit potential and

all other electrodes at zero potential. In this case, all lines of

force are parallel to the imaginary axis and the capacitance per

unit length, Clt between electrodes a/3 and y& is simply

1 logc (a + b) - log, b C| — — 11) 4tt tt = 4i log<( 1+ D

(2)

Similarly, the capacitance pet unit length, C2, between elec- trodes aS and jSy is,* from Fig. l(e)(ii), C'=4-p ,0g<( ,+

«) • • ' • 0)

* Note that eqns. (2) and (3) agree with eqn. (75) of Reference 2.

525
LAMPARD AND CUTKOSKY: SOME RESULTS ON THE CROSS-CAPACITANCES PER UNIT LENGTH OF

, s r

y

b

(C )

0 "'|-— o —"|— b — a'" J*"^*"~

b —J"— °

r

(ii)

y"

r— LDG f b—

]

—LOG£ (a+ b)— —

(

(0

—LOG E a —»| —LDG e

la + b)-

(ii)

Fig. 1.—Transformed cross-sections of general cylindrical capacitor.

(a) Cross-section.

(6) Transformation into j-planc.

(c) Transformation into w-planc. W) Transformation into H-plane.

(i) Orientation I. (ii) Orientation 2. (e) Transformation into v-planc.

(i) Orientation 1 . (ii) Orientation 2.

If we introduce a parameter t = a/b we see that the two

cross-capacitances per unit length for any cylindrical capacitor can always be written in the form

Eliminating r between these two equations, + e 1

• (4)

(5)

• (6)

This identity was given as eqns. (32) and (71) of Reference 2

in connection with cylindrical capacitors of square and circular cross-section, respectively, but its complete generality was not

noticed until some time later (Lampard4 and more recently van der Pauw5

). It follows immediately from eqn. (6) that, if C\ = C2 ,

C, C2 =^logs 2 (7)

It is clear that a sufficient, but not necessary, condition for the equality of C, and C2

is the existence of an axis of symmetry

of the capacitor cross-section, which passes through a pair of

non-adjacent electrode gaps. This was the form in which the theorem was originally stated. It will be clear that the cross- section of any cylindrical capacitor whose cross-capacitances per

unit length are equal can always be mapped into a cross-section having this symmetry property, and hence, in the paper, any such

cylindrical capacitor will be referred to as electrically symmetric.

The non-necessity of the physical symmetry condition has been

pointed out by Thompson. 6 On differentiating eqn. (6), e-4*2C nlC, = - e-^klC, (8)

and, in particular, when C{ = C2 ,

ciCi = - dC2 (9) We shall make use of this result in a later Section to prove some general results concerning the effects of thin dielectric

films.

(3) CONFORMAL MAPPING OF DIELECTRIC

DISTRIBUTIONS

The way in which two simple uniform dielectric distributions behave under the conformal transformations of Section 2 will be examined. Thus consider the physically symmetrical system

of Fig. 2(a). This system was analysed in Reference 2, where

. PERMITTIVITY f

Ii.

(a) .J

PERMITTIVITY (

Fig. 2.—Infinite-plane cylindrical capacitor with dielectric film.

(a) Cross-section.

(/>) Transformation into ;-plane.

526
CYLINDRICAL THREE-TERMINAL CAPACITORS WITH THIN DIELECTRIC FILMS ON THEIR ELECTRODES

rzzzzzizzz

PERMITTIVITY V f (a)

PERMITTIVITY (

PERMITTIVITY e

Fig. 3.—Parallel-plate cylindrical capacitor with dielectric film.

(a) Cross-section.

{b) Transformation into w-plane.

(r) Transformation into z-plane.

it was shown that the cross-capacitance per unit length between

electrodes a/3 and yS is given, when the dielectric thickness is small, by

terms in (^j and higher (10)

If we transform this system [which is to be compared with

that of Fig. lW)(ii)] back into a circular cross-section by the methods of the previous Section, we obtain the system of

Fig. 2(b), where the dielectric is still symmetrically distributed with respect to the capacitor symmetry plane but is no longer of constant thickness. From eqn. (10) it can be seen that for this system each cross- capacitance per unit length is increased by an amount proportional to the square of the dielectric thickness when this thickness is small. Now consider the electrically symmetrical system of Fig. 3(a), which may be compared with Fig. l(e)(i) (with a = b). Trans- forming the system back by the methods of Section 2, we obtain

the physically symmetrical systems of Fig. 2(b) and (c), respectively. It will be seen that the dielectric in the transformed

systems is neither symmetrically disposed with respect to the capacitor symmetry plane nor of constant thickness. A simple calculation on the original system of Fig. 3(a) shows

that, in this case, the cross-capacitance per unit length between

a/9 and yS is

1 + 2

r c 1 + 4 + . . .

. (II)

i.e. the cross-capacitance per unit length between a/3 and yS is increased by an amount directly proportional to the dielectric thickness when the thickness is small. These results suggest that, for small dielectric thickness, the cross-capacitance per unit length of a symmetric cylindrical

capacitor may be written as a Taylor series in terms of the

dielectric thickness, in which the coefficient of the linear term is zero, provided that the dielectric is disposed symmetrically with

respect to the capacitor symmetry plane. That this is so will be demonstrated in Section 5. Up to this point we have been concerned only with the individual cross-capacitances per unit length. It would be of

considerable interest to know what happens to the mean capaci- tance per unit length of a symmetrical cylindrical capacitor when

its electrodes have a thin dielectric film distributed on them

asymmetrically. Thus in the system of Fig. 3 it may be conjectured that the cross-capacitance per unit length between aS

and |3y will be reduced by the same amount, to the first order,

as the increase in the cross-capacitance per unit length between

a/3 and yr>. As will be seen in the next Section, such a result is certainly true for the system discussed there (a slightly simplified version of the system of Fig. 3), and in Section 5 it is shown that

this result is true in general.

(4) THE PARALLEL-PLATE CYLINDRICAL CAPACITOR WITH

A UNIFORM DIELECTRIC FILM ON ONE ELECTRODE

By the methods of Section 2 any cylindrical capacitor can be transformed into an equivalent parallel-plate cylindrical capacitor. When the original capacitor cross-section may be trans- formed into one having symmetry about a line passing through

a pair of non-adjacent electrode gaps, the dimensions of the equivalent parallel-plate capacitor have been shown to be such

527
LAMPARD AND CUTKOSKY: SOME RESULTS ON THE CROSS-CAPACITANCES PER UNIT LENGTH Ol

that each cross-capacitance per unit length is given by C = with the boundary conditions

(1/4772

) log, 2. In this Section we analyse the general parallel-plate cylindrical

capacitor with the additional complication of a dielectric film of constant thickness on one electrode. The capacitor crosssection, together with the co-ordinate system that will be used in the analysis, is shown in Fig. 4.

TO 0

: //!// /-/./ ZZ

PERMITTIVITY (

Fig. 4. Iross-section of parallel-plate cylindrical capacitor with

dielectric film on one electrode only.

The working of this Section will be given in some detail, since the system of Fig. 4 appears to be one of the few general

asymmetric cases in which both cross-capacitances per unit length can be fairly readily evaluated analytically.

(4.1) Cross-Capacitance per Unit Length from a(3 to y5 = C,(e) Electrode aj3 may be placed at unit potential, all other elec- trodes at zero. The lines of force are parallel to the y-axis

and the dielectric boundary is an equipotential. Thus the required capacitance per unit length may be regarded as arising from two capacitors in series:

[CKcfland hence

C,(€) a

A-nb

a

4tt6

- n l-l

4-rr(b -

t (e -

b e

t (€-

f-V (12)

(13)

(14)

In particular, for the symmetrical cylindrical capacitor, a = logt 2 b = TT and substituting eqn. (15) in eqn. (14) and denoting the particular value of C{ by C*,

Cf(6) = (^log.2)

rr e

1) G)'(^)

(16)

(4.2) Cross-Capacitance per Unit Length from otStopy = C2(£) The calculations here are considerably more difficult. It is convenient to take electrode aS at unit potential and all other

electrodes at zero. In region B, with the co-ordinate system

shown, the particular solution of Laplace's equation which

vanishes when y = 0 is + «

Kb(-*> y) = B(k)ejkx sinh kydk . . . (17)

J — 00 and, in region A, is

VA

{x, y) = [/l,(/c>cosh ky + A 2 (k) sinh ky]eJk*dk (18)

J — CD

VA{x, I) = VB

(x, t)

. *y

(19)

(20)

These two boundary conditions, together with Lerch*s theorem on the uniqueness of Fourier transforms, serve to determine the two functions A } (k), A 2(k) in terms of B(k): AJk) = \B(k)(] - e)sinh2A7 (21) A4k) - {B(k)[(\ + e) - (I - 6 )cosh2Af] . (22) On substituting eqns. (21) and (22) in eqn. (IS), + 00 y) = + {(!-€) sinh 2kr cosh ky

-'—00 + [(1 + e) — (1 — e) cosh 2k

} sinh kyB(k)e>k*dk (23)

Setting y = b and using the boundary condition VA

{x, b) = I x > 0

"1 = 0 x <0 \ we obtain, on inverting the Fourier transform of eqn. (18),

}fl(A-){< 1 - e) sinh 2kt cosh kb

(24)

2irl

[(I +c)

-(1

e-Jk*dx .

s) cosh2^f] sinh kb} .... (25)

where B(k) is the unit impulse or Dirac delta function. Using eqn. (26) in eqn. (23) and writing

1 + e

1 - €

(26)

(27)

we find the complete solution for the potential function in region A, namely

.+ »

1 d5) VA(x>y) ==_!_ r r-L

277 J \jk

~* _ ^ + nS(k) e

jkx x

sinh 2kt cosh ky + (m — cosh 2kt) sinh ky^ sinh 2kt cosh kb + (m — cosh 2kt) sinh kb Now, the charge density on the upper electrode is given by

<t(x) = -

*VAx,y)

8^

1>y

1 Jk

(29)

1 + n8(k)^ke'kxF[k, b, t]dk (30)

where

.

1 sinh 2kt sinh kb + (w — cosh 2kt) cosh kb " ' ' '* ~ sinh 2kt cosh kb + (m - cosh 2kt) sinh kb ( '

It seems impossible to evaluate eqn. (30) in closed form, so we

content ourselves with an expansion in powers of /, the dielectric thickness. This procedure is reasonable as we are primarily concerned with the effects of thin dielectric films. Thus,

528
CYLINDRICAL THREE-TERMINAL CAPACITORS WITH THIN DIELECTRIC FILMS ON THEIR ELECTRODES

expanding eqn. (31) in powers of /, and substituting the resulting

expansion in eqn. (30),

[1 + n8(k)

, ., cosh kb „ sinh &6

- + ttS(A:) ih2

Art

l-ttS(A-)

, , .. cosh kb „ sinhJ Aft + terms in / 3 and higher (32) The three integrals occurring here are readily evaluated, for negative x, by contour integration. Results are given in Appendix 8.1. On using these results in eqn. (32) we have,

finally,

a(x) =

1 £

-xtb t (e-l) +

4770 (1 h €

TTX ,, e

itxlb b

\ — eTM/h (J_ £

7rJc/6)2

4 £

~xlb 2er.xlb

1 - (1 - £"*/*) 2

+ (1 _ £

TM/fc)3 J j + terms in (?/6)

3 and higher (33) Here we have reintroduced e by the relation

2 e - 1

1 - w

(34)

The cross-capacitance per unit length, C2 (e), is just the total charge on electrode /3y, namely

C2 (e) = [ dx)dx (35) We now substitute eqn. (33) in eqn. (35), and carry out the

integration, making use of the results given in Appendix 8.2.

1 /t\ /€ — l\ira E~TMlb

T

2 \b)( e ) + +,(?{(—

\

2

- f—\- 4tt2 V6/ V e / 2(_N 6 / (1 -e-TM'*)2 H- £- rta/* + terms in and higher ......... (36)

Of particular interest is the case of the symmetrical cylindrical capacitor for which

a - log. 2 b = 7T .... (37) For this case, denoting the particular value of C2(e) by C*,

£

--a/6 „7Ta £ - 2 —

1 _ g-no/i.

-a/b

cw-^L'-^-G)^)'"- 1 * 2' + terms in ^—^ and higher (38)

We note that the linear terms in eqns. (16) and (38) are equal

in magnitude and opposite in sign, so that the change in the mean capacitance per unit length is only of the second order in the dielectric thickness. Specifically

cfoo 4- cm i = log. 2

e - l\ 2 logs 2

terms and higher (39)

This example thus supports the conjecture that, in the case of a thin dielectric film asymmetrically distributed on the elec- trodes of a cylindrical capacitor, the mean capacitance per unit length remains constant to the first order. Finally let us briefly consider the results for the general

parallel-plate cylindrical capacitor with no dielectric film. Setting t = 0 in eqns. (14) and (36) we have (on dropping the arguments of C, and C2 )

Cl

4tt6

477- lOgc ( I - F-TMlb

)

(40)

(41)

As a partial check on the fairly complicated calculations of

this Section we note that the introduction of a parameter — gTia/b

J (42)

enables eqns. (40) and (41) to be written in the standard parametric form of eqns. (4) and (5).

(5) GENERAL RESULTS CONCERNING DIELECTRIC

FILMS IN CYLINDRICAL CAPACITORS

In this Section we give proofs of the following general results, which have been suggested by the working of Sections 3 and 4. (a) In an electrically symmetric cylindrical capacitor, an

arbitrary thin dielectric film on the capacitor electrodes produces

only a second-order change in the mean capacitance per unit

length.

{b) In a physically symmetric cylindrical capacitor, a thin

dielectric film on the capacitor electrodes, which is disposed

symmetrically with respect to the capacitor symmetry plane, produces only a second-order change in each cross-capacitance

per unit length. Before proceeding to the proof of these results we establish a preliminary lemma. This lemma states that, given any

cylindrical capacitor with a thin dielectric film on its electrodes, there exists an equivalent cylindrical capacitor containing no

dielectric,* whose electrodes lie wholly within the space occupied

by the dielectric film, such that the electric fields, at all points

in the dielectricless region of the two capacitors, are equal to the first order. The significance of this lemma is that it shows, for dielectric films whose thickness is small, that the perturbation required to obtain the equivalent electrode system is both small, of the same order as the dielectric thickness, being in fact bounded by

the dielectric itself, and independent of the field configuration,

i.e. independent of the individual electrode potentials. Thus consider the cross-section of an arbitrary cylindrical capacitor and, in particular, a small region near electrode a/3, say (assumed at zero potential without loss of generality), as shown in Fig. 5.

• Throughout the paper the term dielectric is used in the sense of a material

substance.

529
LAMPAR D AND CUTKOSKY: SOME RESULTS ON THE CROSS-CAPACITANCES PER UNIT LENGTH OF

ENLARGEMENT OF C

ELECTRODE

PERMITTIVITY f EA

(s)

DIELECTRIC BOUNDARY

Fig. 5.—Cross-section of general cylindrical capacitor with arbitrary thin dielectric film.

In the following discussion we assume that:

(i) A(5) is small compared with the length of the line of force terminating at 5.

(ii) A'(s) is such that the field is almost normal to the dielectric boundary.

With these assumptions, the field EB(s) in the dielectric can

be regarded as essentially constant for a given .y and. on equating normal components of electric displacement,

EA(s) cos BA = eEB

(s) cos dB

i.e. EB

(s)

Using the identity

tan2 9B

e tan 8A and noting that for small A'(.y)

tan 9B = A'(j)

1 + tan 2 6A

tan d R

we haveEB(s) = -EA

(s)

(43)

(44)

(45)

(46)

(47)

• =-EA(s){\ + ofA20?)]} (48)

Thus for small A(a) the potential function at the dielectric boundary is

V(s) = 1a(5)£4 (5) + o[A2

(.y)] . (49) Now consider the same region of the capacitor but with the

dielectric removed and the electrode surface at j advanced a

distance xA(s) towards the former position of the dielectric boundary. The potential at the dielectric boundary is now

simply

V*(s) = (1 - x)Ms)E*(s) + o[A2 (*)] . . (50) where asterisks have been used to denote the values of V(s) and EMs) in this situation. Clearly, if V(s) = V*(s) for all s, then so must the electric

fields in region A also be equal; in particular we must have EA

(s) = E*{s) for all s. Equating the two expressions for Vis) and solving for x, x = —— 4- o[A(.?)] . . . . (51)

which result is independent of s to the first order and also independent of the way in which the field arises. Thus we have shown that the electric field in the region

unoccupied by dielectric remains constant to the first order, if the dielectric and electrode systems are replaced by a perturbed

electrode system (without dielectric) such that each new electrode element is a distance A(s)(e — l)/e from the corresponding

original electrode position. Since (e — l)/e < 1, the perturbed

electrode system of the equivalent dielectricless capacitor lies wholly within the region formerly occupied by the dielectric, which proves the lemma.

Finally, the capacitance per unit length between electrodes a/3 and yS, say, can be computed by assuming that electrode y8 is at unit potential and all other electrodes are at zero potential. Hence

C, = a(s)c/s (52)

where o(s), the charge density per unit length, may be written

in terms of the electric displacement at the electrode surface. To the first order, this electric displacement is identical with the normal component of the electric field at the dielectric, and

hence

<j(s) = ^EA (s){\ + 1"^ 2J"

,/2 + 0 [A(*)] (53)

= ~f4

(.t) + o[AW] (54) We have already seen that the electric field £a(j) remains

constant to the first order in the equivalent dielectricless capacitor, and so, on using eqn. (54) in eqn. (52), it is clear that the same conclusion applies to the capacitance per unit

length, C|, and by similar working to the other cross-capaci- tance per unit length, C2

. As an example of the use of this lemma we may compute the

first-order changes in the cross-capacitances per unit length fcr the system of Fig. 4 due to the presence of the dielectric film. Then, according to the lemma, the equivalent dielectricless capacitor would have the same disposition, but the distance b between the electrodes must now be reduced by an amount

r(e — 1)/e. Thus, using eqns. (40) and (41),

C,(€) to the first order . (55)

C2(e) = —7-^1 log,

47T"

I — exp / —

h-tC,(e)

C2

(e) = -

(57)

to the first order (56) On expanding eqns. (55) and (56) in powers of /(e — l)/e,

/ t e - 1

1

1 / e — 1 -na 4TT2 b ~T

log, (1

higher termsj

alb)

p — rzafb- ,r + higher terms (58) b 1 - £--"/*

which of course agree with eqns. (14) and (36), to the first order.

530
CYLINDRICAL THREE-TERMINAL CAPACITORS WITH THIN DIELECTRIC FILMS ON THEIR ELECTRODES

To prove the first general result, mentioned at the beginning

of this Section, consider an electrically symmetric cylindrical capacitor with an arbitrary thin dielectric film on all its elec- trodes. By the lemma and its subsequent discussion it follows

that there is an equivalent dielectricless capacitor, with the same cross-capacitances per unit length to the first order, but whose electrodes lie wholly within the region formerly occupied

by dielectric. The change in the mean capacitance per unit

length, CM, is then given by

sc. = i(</C, + dC2 ) + terms of higher degree

(59)

(60)

But for the electrically symmetric cylindrical capacitor

(C, = C2) we have, from eqn. (9),

and so

dCi = - dC2 (if C, = C2) . 8CM = 0 (to the first order)

(61)

(62) which proves the first result. To prove the second general result, consider a physically

symmetric cylindrical capacitor C with a thin dielectric film on

all its electrodes, which is disposed symmetrically with respect

to the capacitor symmetry plane. The dielectric in such a

are both (

I /4-n-2 ) log, 2 and thus remain constant to the first order, which proves the second general result.

(6) CONCLUSION

The application of conformal-transformation methods to certain types of cylindrical 3-terminal capacitors, whose elec- trodes carry thin dielectric films, has suggested some general

results concerning the effects on the cross-capacitances of such

dielectric films. The detailed computation of the cross-capacitances of a parallel-plate cylindrical capacitor (with a thin, highly asymmetric dielectric film on its electrodes) has further substantiated these conjectures, and proofs are given in the final Section.

It should be realized that the results of the paper apply only

to dielectric films which maintain a constant cross-section throughout the length of the capacitor. More generally,

dielectric films will be distributed in a 3-dimensional pattern over the electrode surfaces. In this case the potential problem

is correspondingly 3-dimensional and hence the powerful methods of conformal transformation are no longer available

for its solution. Such problems require further study.

(7) REFERENCES

(1) Thompson, A. M., and Lampard, D. G.: 'A New Theorem

in Electrostatics and its Application to Calculable Stan- dards of Capacitance'. Nature, 1956, 177, p. 888.

C

H ft

s

Fig. 6.—Sequential placement of dielectric film in a physically symmetric cylindrical capacitor.

capacitor may be considered to be introduced in two steps as shown in Fig. 6. When we consider the equivalent dielectricless capacitors,

which exist by the lemma above, we have, on making use of

eqn. (9),

SC,' = - SC2 (to first order) .... (63)

SCJ' = - 8C2 (to first order) .... (64)

Also, because of the symmetry of the dielectric film,

SC,' = 8C{

SC2 sc;-

(65)

(66) On superposing the two sets of equations to obtain results for the whole capacitance, C,

SC, = SC,' + SC,' - 0 (to first order) . . (67) 8C2 = SC2 + SC2 = 0 (to first order) . . (68)

which proves the second general result, namely that in this case

the individual cross-capacitances per unit length remain constant

to the first order. Alternatively we may note that, by the lemma, the equivalent

dielectricless capacitor is also physically symmetrical, and hence

to the first order the individual cross-capacitances per unit length

(2) Lampard, D. G.: 'A New Theorem in Electrostatics with

Applications to Calculable Standards of Capacitance',

Proceedings I.E.E., Monograph No. 216 M, January, 1957

(104 C, p. 271).

(3) Copson, E. T. : 'Theory of Functions of a Complex

Variable' (Oxford, 1948).

(4) Lampard, D. G.: C.S.I. R.O. Division of Electrotechnology,

Internal Report, March, 1957.

(5) van der Pauw, L. J.: 'A Method of Measuring Specific

Resistivity and Hall Effects of Discs of Arbitrary Shape',

Philips Research Reports, 1958, 13, p. 1. (6) Thompson, A. M.: 'The Cylindrical Cross-Capacitor as a Calculable Standard', Proceedings I.E.E., Paper No.

2887 M, May, 1959 (106 B, p. 307).

(8) APPENDIX

(8. 1 ) Values of Integrals occurring in Eqn. (32) The integrals occurring in eqn. (32) may be evaluated (for negative x) by summing residues at the poles k = — jmrfb

with n = 0, 1,2... The results are: „+«>

I —. H- ttS(A)

,jk

, ., coshAA „ ke'kx -r-7—ridk

sinn kb

(69)

1 4^6 1 - £-*/* (70)

531
LAiMPARD AND CUTKOSKY: DIELECTRIC FILMS IN CYLINDRICAL CAPACITORS

1 8^ k

l e'kx -dk

1 Airb

sinh2 kb

4nb2

TTX ,.

1 _ e

r.xlb b

E

-x/b ^ (1 — e-x/b)2 (72)

1 87T2 + nSik)

fob3 „5i

1

8ffl!> 3

, , .. cosh £6 „ smrr kb

/mix\2

, /Trnx\ M +4 (t-) +2

L-TTX/b

+

(1 _ £TM/*)3 (74)

(8.2) Values of Integrals occurring in Eqn. (35) When carrying out the integration of eqn. (35), with o(x) given

by eqn. (33), the following results are needed:

— a

1 - e

5"/* — OO

TTX ,. b

(1 _ £-*/(-)

—^rfr = log, ( I - e -««/*)

rrf* = - - TM -A b

1 _ e--o/i lOg; (1 - £-"°/ft )

(75)

(76)

(1 - t--'/*) 3

? c--ab

h + -—rrz^ - 2 log, (1 - e-TM*)

(1 _e-TM°/*)2

(1 -£--<>/*) (77)

Printed ut Great Britain by UNWIN BROTHERS LIMITED, WOKING AND LONDON

532
A COMBINED TRANSFORMER BRIDGE FOR PRECISE COMPARISON OF INDUCTANCE WITH

CAPACITANCE, Andrzej Muciek, IEEE Trans. Instrum. Meas. IM-32 , September

1983.

A combined transformer bridge, composed of two alternately balanced

transformer bridges, designed for precise comparison of inductance with

capacitance standards has been analyzed. A number of possible sources of error have been studied and experimentally examined in a constructed model of

the combined bridge. The experimental results confirmed an ability to measure inductance in reference to capacitance with an accuracy up to 10

parts per million (ppm) in the frequency band 0.16-2 kHz.

AC/DC TRANSFER STANDARDS AND INSTRUMENTS

A Dual-Channel Comparator for AC-DC Difference

Measurements

Earl S. Williams and Joseph R. Kinard (1984) 537

An Automatic System for ac/dc Calibration

K. J. Lentner and Donald R. Flach (1983) ..... 542

A..C. Voltage Calibrations for the 0.1 Hz to 10 Hz Frequency

Range

Howard K. Schoenwetter (1983) . 548

The Practical Uses of ac-dc Transfer Instruments

Earl S. Williams (1982) 598

A Thermoelement Comparator for Automatic ac-dc

Difference Measurements

Earl S. Williams (1980) ........ 631

An International Comparison of Thermal Converters

as ac-dc Transfer Standards

0. P. Galakhova, S. Harkness, Francis L. Hermach,

H. Hirayama, P. Martin, T. H. Rozdestvenskaya, and

Earl S. Williams (1 980) 636

A Semiautomatic System for ac/dc Difference Calibration

Keith J. Lentner, Clifton B. Childers, and

Susan G. Tremaine (1980) 640

An RMS Digital Voltmeter/Calibrator for Very-Low

Frequencies

Howard K. Schoenwetter (1978) 646

A Fast Response Low-Frequency Voltmeter

Bruce F. Field (1 978) 656

AC-DC Comparators for Audio-Frequency Current and

Voltage Measurement of High Accuracy

Francis L. Hermach (1976) 661

Thermal Current Converters for Accurate ac Current

Measurements

Earl S. Williams (1976) 667

An Investigation of Multi junction Thermal Converters

Francis L. Hermach and Donald R. Flach (1976) ......... 672

A Low-Temperature Direct-Current Comparator Bridge

D. B. Sullivan and Ronald F. Dziuba (1974) 677

535
Thermal Voltage Converters and Comparator for Very

Accurate ac Voltage Measurements

E. S. Williams (1971 ) 682

Design Features of a Precision ac-dc Converter

Louis A. Marzetta and Donald R. Flach (1969) 692

*

536
A Dual-Channel Automated Comparator for AC-DC

Difference Measurements

EARL S. WILLIAMS and JOSEPH R. KINARD, senior member, ieee

Abstract-An automated ac-dc difference calibration system is described. The system incorporates a new electronic comparator which

determines ac-dc differences of thermal voltage converters (TVC's) by simultaneously measuring the difference between nearly equal ac and

dc voltages with both the test and standard instruments. The comparator consists essentially of two practically identical channels each containing a digital-to-analog converter balancing circuit, an operational

amplifier to amplify voltage imbalance, and an integrator circuit in which

a capacitor is charged during an accurately controlled 10-s period. The

difference between the ac and dc voltages applied to the test and stan- dard TVC's is computed from differences in capacitor voltages, and the ac-dc difference is derived from the variation between the test and stan- dard channel indications of this voltage difference. Measurements are made in about half the time required for our manual procedures.

I. Introduction

AC AND DC voltages are compared using a thermal voltage

. converter (TVC) which responds nearly equally to ac and

dc voltages. Those discussed here are electrothermic. Typically,

they consist of ranging resistors and a thermoelement—a vac- uum bulb containing a heater wire with a thermocouple at- tached to its midpoint. The output, E, of the thermocouple

is usually 7-12 mV with rated heater current. The TVC is

Manuscript received August 23, 1984. This work was supported by

the Calibration Coordination Group of the U.S. Department of Defense.

J. R. Kinard is with the Electricity Division, National Bureau of Standards, Gaithersburg, MD 20899.

E. S. Williams (retired) was with the Electricity Division, National Bureau of Standards, Gaithersburg, MD 20899.

connected alternately to an ac voltage, Va , and to both polarities of a dc reference voltage, Vd . Va

is measured relative to Vd by observing the output EMF, E, of the TVC and changes

in output, Ea -Ed , where Ea

is observed with Va applied, and

Edis observed with Vd applied. When the difference between Va and Vd is small, the proportional difference can be computed from (1):

(Va - Vd) _

(Eg - Ed)

7/

, , m c— + 6 - v) Vd nE

The factor « relates small changes in TVC heater current to corresponding changes in output EMF, E. It is near 2.0 (a square-law response) at very low heater currents, but is usually 1.7-1 .9 at rated heater current. 5 is the ac-dc difference of the TVC. It is negligible at low frequencies, but becomes

significant at frequencies of 20 kHz and higher in most TVC's. The ac-dc difference correction for a TVC is determined by

comparing it to a similar standard instrument whose correction

,

bs

, is known [1 ] . Ac and dc voltages are applied to both TVC's

in parallel, and in most procedures, the ac and dc voltages are adjusted for equal outputs from the test TVC, i.e., Ea = Ed

(1). The test instrument thus indicates that Va = Vd . The actual proportional difference between the ac and dc voltages is then measured by observing E and changes in E from the stan- dard TVC. The correction to the test TVC, 6 r

, is computed

using an equation similar to (1):

537
WILLIAMS AND KINARD: DUAL-CHANNEL AUTOMATED CONVERTER

IEEE 488

DC SOURCE

AC SOURCE

DUAL CHANNEL COMPARATOR 1

IEEE 488

~7\

STO. TVC

CONTROL FROM

IEEE 488 o-i RELAY —

9

. FREQ. ~7 COUNTER

VOLTAGE

NULLINO DAC 7^

IEEE 4S8

INTEGRATOR

DVM

TE3T

TVC

^ 1- VOLTAGE

NULLING DAC

AMP/> ' \—

IEEE 488 L

INTEGRATOR

IEEE 488 V

CONTROLLER

PRINTER PLOTTER >

Fig. 1. Block diagram of automated ac-dc difference calibration system.

6 t = (Ea

- Ed )/nE + 8s (2) where subscripts t and s refer to test and standard. With a manual comparator, the precise adjustments of ac and

dc voltages for equal outputs from the test TVC are tedious and

a skilled operator is necessary [2] . These adjustments have been made automatically but the process is slowed significantly

[3] . In the alternative procedure used in the new comparator,

data is collected more quickly because the ac and dc voltages

are set nearly equal and not further adjusted during one deter- mination. The ac voltage is trimmed automatically after each

determination to compensate for drift and to avoid the dif- ficulty, discussed below, of a large difference between ac and

dc applied voltages. In this new approach the difference be- tween the unequal ac and dc voltages is measured by both the

test and standard TVC's, and, therefore,

(Va ' Vd ) _ (Eas

- Eds ) _

(Eat - Edt ) , f + o<. ———+ 6 t

. ns {Es ) n t {Et ) (3)

Eliminating the first expression and solving the remaining two

for 8 f

, we have

5r =

(Ea s Eds ) (Ea t Edt)

ns (Es ) n t (Et ) (4)

II. Automated System

The entire automated ac-dc difference calibration system is shown in block diagram form in Fig. 1 . A photograph of the equipment appears in Fig. 2. The system contains programmable ac and dc voltage sources, a high -voltage relay for switching between ac and dc, voltmeters and frequency counters, a desktop computer to provide IEEE-488 bus control, and the two-channel comparator itself. The two-channel comparator

circuit is sketched in Fig. 3. It uses elements from a commercial

data acquisition system (DAS) including programmable digital- to-analog converters (DAC's), programmable relays (Kl to K6

and K9), and an auto-ranging voltmeter (VM). An additional board was assembled containing the DAC balancing circuits, amplifiers Al to A4 and their related resistors and capacitors.

The DAC's supply up to 10 volts to Lindeck-type circuits with 5-k£2 total circuit resistance. With only one range, the

12-bit resolution is insufficient for an appropriate null. A twopole four-position input switch allows the operator to select either a 5- or 10-H resistance in the divider circuit, thus pro- viding either a 10- or 20-mV range. The measurement procedure is as follows. The operator provides ranges, voltages, frequencies, and record-keeping information to the controller interactively. The ac and dc supplies are then programmed to the appropriate values, and dc is applied

to the TVC's. With the DAC set to zero and K3 closed, the standard TVC output, Eds , is measured by the voltmeter, and

a DAC voltage is computed and supplied to null the input to

amplifier Al . The output of Al , with gain of 10 000, is then measured by closing relay K2, and the DAC signal is trimmed

in several steps until the amplifier output is less than 20 mV.

The same balancing procedure is followed in the test channel. When both balances have been completed, ns (Es ) and nt (Et ) are computed and printed along with other test parameters. 538
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

Fig. 2. Automated ac-dc difference calibration system.

K3/

I V? ..."2 *2

0o 1 STI R3 HPConntctor

Fig. 3. Dual-channel ac-dc difference comparator.

Voltages are then applied to both TVC's in the test sequence

(ac, dc+, dc-, ac), and after each one has stabilized, the relay K7, with a contact in each channel, is closed to charge the two

capacitors during an accurately measured time interval of 10 s. The capacitor voltages, four from each channel for a single measurement, are then measured with the voltmeter by closing

relays Kl and K4 in turn, after which the capacitors are dis- charged by closing two-pole relay K8. Providing they are stable over the time required for one determination, amplifier offsets cancel because the results are collected in paired sums and

differences. For the standard channel, the relation between the voltage, ein , at the input to Al and the voltage , eout , on the integrator

capacitor is

enut - R3C1 - f

J

Rl + R2

Rl

dt = -ein X 10 s (5)

with Rl = 10 £2, R2 = 100 kfi, R3 = 1 Mft, CI = 1 juF, and an

integration time of 10 s. Nominally equal values of resistance and capacitance are used in the standard and test channels, so the same nominal relations apply. If the eight voltage readings

are reversed in sign and set equal to e2 , e4 , e6 , and es in the standard channel, and e3 , e5 , e 7 , and eg in the test channel, then using (5), (4) may be written as

5 X (e 2 - e4 - e6 + e8 ) 5 X (e 3 - es - e1 + e9 ) 106 n t Et -106 + 5<

(6) The numerator in (6) gives twice the actual change in EMF, so the constants are adjusted appropriately. These constants are

also adjusted slightly to compensate for departures from nominal in the integrators and amplifiers. The test TVC correction

is computed using (6), with the appropriate ac -dc difference correction, 5^, determined from an equation devised to simulate

the standard TVC correction curve. Since this two -channel method requires the knowledge of n for both test and standard, any uncertainty in either n contributes to uncertainty in the final ac-dc difference. Should

either or both terms in (4) become large due to inequality be- tween applied ac and dc voltages, the calculation contains the

difference of two large terms each involving an n. Since the n

value can only be determined to 1 or 2 percent, a constraint is added to system that the 1st term in (4) be 50 ppm or less to

limit the uncertainty contributed by n to 1 ppm, or less. With

this constraint the procedure places an upper limit of about 50 ppm on the difference between the applied ac and dc voltages. Equally satisfactory results are obtained by a "two-voltmeter"

539
WILLIAMS AND KINARD: DUAL-CHANNEL AUTOMATED CONVERTER

IBBV zE

[0 Ik 10k FREQUENCY (Hi)

100k

Fig. 4. Voltage and frequency limits of calibration system.

method similar to that used by Inglis in a semi-automatic sys- tem [4] . Instead of the analog integrators, an external digital voltmeter connected to relay K9, and the voltmeter located in the DAS, are used to measure the amplified voltage imbalances.

After the DAC balances are made, relays K2 and K9 are closed and remain closed while the ac and dc voltages are applied. The

voltmeters are read alternately and the two sums are accumulated. If ten readings are summed for each channel with the measurement sequence ac, dc+, dc-, ac, and the sums set equal to e2 through e 8 and e 3 through e9 , then 5, can still be

obtained from (6) with no change.

III. Other Measurements

Measurements of the response characteristic, n, and comparisons of ac and dc voltages are two types of measurements made

using only one channel of the comparator. For these cases, each EMF measurement is the average of 10 voltmeter readings taken in about 10 s.

A. Measurements of n The factor n is ordinarily measured at 10 percent intervals from 50 to 110 percent of rated voltage. For instruments to be calibrated at rated voltage, however, n is usually determined

at 80, 90, and 100 percent. In the middle voltage range, small accurate voltage steps are easily made and the stability of the system is optimum. Tests are made, therefore, most conveniently on the 100-V range of multirange TVC's, and the re- sulting n values are usable with any range when related to the TVC output, E. Low-voltage single-range TVC's are usually

tested with additional resistance in series, so that 50-100 V

may be applied. The output of the TVC is connected to the "STD" channel, and the operator enters the lowest and highest

voltages applied. After the TVC has stabilized at the measure- ment voltage, the DAC balance is made as described above, and

relay K2 is closed connecting the amplifier to the voltmeter. The TVC input is increased and decreased alternately by 0.1 V,

A V, and the resulting changes in TVC output are computed

from voltmeter readings.

With only dc applied, the relationship between small changes

in input voltage, AV, and changes in the output voltage, AE , can be given by rewriting (1) as AV/V=AE/nE or n = (AE/E)/(AV/V). (7)

As in the two-voltmeter method, 10 voltmeter readings of the output are averaged to obtain one measured value. One AE

is calculated from measurements taken in the sequence {V,

V + AV, V). The last reading with V applied from the previous

set is then taken as the first reading in the next set. Four values of n are computed from four values of A £ requiring nine volt- meter measurements. These data are then used to determine

constants for a straight line from which the appropriate value of n can then be found for any value of E.

B. Comparison ofAC and DC Voltages

AC voltage, such as from an ac calibrator, can be measured

relative to a dc -reference voltage by a procedure similar to that used for the determination of n. The system applies dc+, ac, dc-, to a TVC in that order, and its output is monitored by

the "STD" channel with the DAC balance and voltmeter readings performed as described above. Computations are made

using (1) with (Ea

- Ed ) computed from three voltage readingse x ,e2 ,e3 —%o that

2(Ea -Ed ) = e l +2e2 - e 3

where e t and e3 correspond to the two directions of dc volt- age, and e2 to the ac voltage. Again, four values of EMF dif- ference are computed from nine voltages— each being the aver- age of 10 voltmeter readings. When low voltages are being

compared, the voltage drops along the leads from the sources

to the TVC may become significant. Dedicated cables are provide with total inner and outer resistance equal for both ac and

dc. These connections also permit voltage sensing of the dc if required.

IV. Results

A significant reduction in the time required to perform routine calibrations is achieved with this system. Four determinations, for one set of test conditions, are made and averaged in about half the time required with our manual procedures. The

voltage and frequency operating ranges, shown in Fig. 4, are determined primarily by the availability and cost of a program- mable ac voltage source. The region between 100 kHz and 1 MHz represents a maximum volt -Hertz product of 10 7

. Test

540
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 198S

results taken over the operational region rarely differ by more

than 10 ppm from data obtained manually. The standard devia- tion is generally less than 5 ppm in the middle -voltage and middle-frequency region indicated approximately by the cross- hatched region in Fig. 4. For well-constructed single-range

TVC's, standard deviations of 2 ppm or less are achieved. At

extreme voltages and frequencies standard deviations are usually less than 10 ppm. Although this system was built as a re- search tool for investigation of new thermal voltage conversion techniques and instruments, it has been used for some routine

calibrations. Another system, of very similar design and dedicated to routine calibration work, is under construction. These

automatic systems are expected to fulfill most of our TVC cali- bration requirements.

AcKNOWLEDGMENT

The authors would like to thank F. L. Hermach for his helpful suggestions.

References

[1] F. L. Hermach and E. S. Williams, "Thermal converters for audio- frequency measurements of high accuracy," IEEE Trans. Instrum.

Meas., vol. IM-15, pp. 260-268, Dec. 1966.

[2] E. S. Williams, "Thermal voltage converters and comparator for very accurate ac voltage measurements," /. Res. Nat. Bur. Stand.,

vol. 75C,pp. 145-154,1971.

[3] —, "A thermoelement comparator for automatic ac-dc difference measurements," IEEE Trans. Instrum. Meas., vol. IM-29, pp. 405-

409, Dec. 1980.

[4] B. D. Inglis, "Evaluation of ac-dc transfer errors for thermal converter-multiplier combinations," Metrologia, vol. 16, pp. 177-

181, 1980.

541
Il l I TRANSACTIONS ON INSTRUMENTATION AND MliASURKMKNT. VOL IM-32, NO I. MARCH 1983 An Automatic System for AC/DC Calibration

K. J. LENTNER, MEMBER, IEEE, AND DONALD R. FLACH, MEMBER, IEEE

Abstract—An automatic ac/dc difference calibration system using

direct measurement of thermoelement EMFs is described. The system

operates over a frequency range from 20 Hz to 100 kHz, covering the

voltage range from 0.5 V to 1 kV. For all voltages, the total uncertainty

(including the uncertainty of the specific reference thermal converters

used) is 50 ppm at frequencies from 20 Hz to 20 kHz, inclusive, and

!00 ppm at higher frequencies up to 100 kHz. In addition to ac/dc

difference testing, the system can he used to measure some important

characteristics of thermoelements, as well as to calibrate ac voltage

calibrators and precision voltmeters. Results of intercomparisons between the new system and the manual NBS calibration system, using

single-range, coaxial-type, thermal voltage converters as transfer standards, are reported. The results indicate that the ac/dc differences measured are accurate to well within the combined total uncertainty

limits of the two systems.

Introduction

TNTIL RECENTLY, techniques for performing preci-

\_/ sion ac/dc difference measurements generally made

use of manual testing methods which utilize photocell

preamplifiers and light-beam galvanometers as voltage detectors, in conjunction with manually balanced comparators

[

1 ], [2]. Careful attention to the reduction of systematic and

random uncertainties in measurement methods, thermoele- ments (TEs), and thermal voltage converters (TVCs) has re- sulted in sufficient confidence in test data to allow results to be reported with total uncertainties at the 10-100-ppm level (or better, in special cases) over wide voltage and frequency

ranges [3], [4]. These manual test methods, however, are very time consuming and subject to errors due to operator fatigue

and lack of skill. More recent work [5], [6] overcame some of

the deficiencies of manual lest methods. However, the ref- erenced systems still require the use of a TE voltage comparator which is not available commercially. Because of the need

to custom build it, the comparator adds to a system's cost and

complexity.

Stringent requirements arc placed on the measurement

system to resolve low-level voltages (about 10 nV), and to overcome the effects of large temperature coefficients and dc

reversal differences of the TEs, as well as instabilities of the voltage sources. However, with the advent of stable and ac- curate analog voltage amplification techniques in combination

with digital electronics, digital voltmeters arc available with

stability, linearity, resolution, and accuracy sufficient to meet

the requirements of automated ac/dc difference measureManuscript received September 24. 19X2. This work was partially lunded

by the Calibration Coordination Group of the U.S. Department of Defense. The authors are with the Elcctrosystcms Division. National Bureau of Standards, Washingion. DC 20234.

ments. Programmable ac and dc voltage sources which lend themselves to adaption for automatic testing are also available. Advances in analog/digital interfaces have improved the ease with which computers can be used to control these instru- ments. The system to be described in this paper was designed to overcome the deficiencies of present test methods by taking

advantage of the technological advances described above.

Specifically, some objectives were to: 1) utilize a desk-top

computer; 2) use commercially available programmable in- struments whenever possible; 3) eliminate as much operator

intervention as possible in order that testing could be done

completely automatically at one fixed voltage range of the test unit; 4) eliminate the additional cost and complexity of using

a TE voltage comparator; 5) use test procedures which assure

that the test TE's output EMF is held as constant as possible; 6) eliminate the effects of operator fatigue and lack of skill; and 7) achieve about the same level of uncertainty as is pres- ently possible for general purpose ac/dc difference manual

testing methods. The system operates over a frequency range from 20 Hz to 100 kHz, covering the voltage range from 0.5 V to 1 kV. For

all voltages, the total uncertainty (including the uncertainty

of the reference TVCs used) is 50 ppm at frequencies from 20 Hz to 20 kHz, inclusive, and 100 ppm at higher frequencies up to 100 kHz. In addition to performing ac/dc difference

tests, the system can be used to calibrate ac voltage calibrators or precision voltmeters. Furthermore, as will be discussed later, automatic measurements of a TE's dc reversal difference, re- sponse time, and the value of the exponent in the expression relating a TE's heater current and output EMF can also be made.

System Hardware

The new system differs from most present techniques for ac/dc difference measurements in that no TE voltage comparator is used. Eliminating the comparator places additional requirements upon the stability of the system's ac and dc

voltage sources, the digital voltmeter (DVM) which measures

a TE's output EMF, and the switches which select a voltage

to be measured by the DVM. The hardware development

problem of obtaining low-thcrmal-EM F switches which could be automatically controlled was solved during previous work

[5], Manually operated low-thcrmal-EM F switches were

modified by linking them to solenoids with 24-V operating coils energized by means of programmable relays. TE voltages of about 10 mV must be measured to within about 10 nV to achieve 1 -ppm comparison precision, and the

542
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT. VOL. IM-32, NO. !, MARCH 1983

DESK TOP COMPUTER

(OTC)

1 I/O SLOTS

I/O EXPANDER

SLOTS

1 2 3 4 5 6 7

_3 p*1( BIT1 PARALLEL

REAL

TIME CLOCK

EL

DC SOURCE

I

RELAY AC

ACTUATOR SOURCE

DC +

-:

*t- DC- • f

1

\

100/1 ATTENUATOR

: ' VOLTAGE h :

STD(^E, LOW-THERMAL-EMF SWITCHES ZZ3tE, 5) UUT

__DIGITAL DATA t CONTROL LINES ANALOG SIGNAL LINES

I I

I I

Fig. I. System block diagram.

DVM used to measure these voltages has the resolution nec- essary to make such a voltage measurement when extensive averaging of data is used. The DVM has an input resistance of about 10 GQ, a 60-Hz common-mode noise rejection >140

dB, and a dc common-mode noise rejection > 1 50 dB. On its most sensitive (10-mV) range, the peak-to-peak low-frequency

(dc to about lC Hz) noise of the DVM, with its input shorted, was measured to be a'ucut 50 nV. The dc voltage source has lO-V, 100-V, and l-kV ranges

with six-digit voltage setting capability on each range. Its specified accuracy is 50 ppm of setting to 10 V, and 1 00 ppm

above 10 V. The ac voltage source also has six-digit voltage

setting capability for voltage ranges of l V, 10 V, 100 V, and

l kV, with specified accuracies in the range of about 0.02 to

0.1 percent depending on voltage and frequency ranges. Some

typical results of stability tests of these sources will be discussed

later. A block diagram of the system is shown in Fig. I. The

desk-top computer serves as the system controller and has about 60 kbytes of user-available memory. Magnetic tape

cartridges are used for program and data storage. A video

display terminal (CRT) is used for data display during testing

and also as an aid during software development. Since the computer only has three input/output slots, an I/O expander

is needed to accomodate the required instrumentation. The

real-time clock is used for precise control of the time interval between applications of ac or dc voltages during a test, and the

printer and plotter provide test records. 543

As indicated in Fig. I, communication between the computer and the stimulus and measurement instrumentation is via the lEEE-488 bus. The digital panel meters (DPM's)

monitor the voltage outputs from the dc and ac sources. During

testing, readings of both the dc and ac source's output voltages are obtained by the DPMs. These voltage readings are made

before the voltages are applied to the test circuit. If the measured voltages differ by more than 0.5 percent from the pro- grammed values, the test is automatically aborted. High ac- curacy in these voltage readings is not necessary, since TVCs

can readily withstand an applied voltage which is 1 20 percent

of rated value. For the ac source, the output voltage is atten- uated by a factor of 1 00 before being connected to the DPM.

The attenuation is necessary in order to overcome the volt- age-frequency product limitation of the DPM. The counter shown in Fig. I monitors the frequency of the ac source. If the counter reading differs by more than about 10 percent from

the programmed frequency, the test is automatically aborted. The relay actuator is used to apply 24-V dc to the operating

coils of the high-voltage relays as well as to the coils of the

solenoids which operate the low-thermal-EMF switches. These

switches select the output voltage of the standard TE (Es ) or

the unit-under-test TE (E, ). The voltages, Es and are applied directly to the input of the DVM, and the DVM voltage

readings are then stored in the computer's memory for later processing.

AC/DC Test Procedure

One of the advantages of using a TE voltage comparator is that, since small differences between the standard and test TE

EMFs are measured directly, the output EMF of the test TE

can be allowed to drift slightly from its reference EMF value when ac, +dc, or —dc voltages are applied [2], [6]. In this new

system, since no TE voltage comparator is used, the require- ment for holding the test TE's EMF constant is quite stringent.

However, with automation and the rapid response time of the voltage sources (typically 50 tts for the dc source and a few

milliseconds for small voltage changes for the ac source), the constant output requirement can be readily satisfied. The

techniques used will be discussed in this section. For a TVC with response defined by E = KV", where E is the output EM F of the TE in the TVC, V is the voltage applied,

and K and n are parameters characteristic of the particular

TVC, the ac/dc difference is defined as

5 =

Vg ~ V_d Va

(I)

where Va is the applied ac voltage, and is the average of the two polarities of dc voltage; i.e.,

Va =

V+ + V- (2)

Va , V+, and V- are voltage values which give the same output

EMF of the TE. In a manual test system using a TE voltage

comparator, the voltage difference Va — Va is determined by

a second TVC, the standard. If the standard's response is defined as
LENTER AND FLACH AUTOMATIC SYSTEM FOR AC/ DC CALIBRATION

Es = KS VS

' (3)

it can be shown [2] that the ac/dc difference of the test TVC

is

5 Esa Esj r dEs

. 0, = h 0 V = — h 5 f s^sd is £. (4)

where £sa is the standard TE's EMF when the voltage Va

is applied, Esa

- is the standard TE's average EMF when voltages

V+ and V- are applied (cf. (2)), ns is the standard's dimensionless characteristic n, and 55 is the ac/dc difference (in parts per million) of the standard determined as described in [

1 ], [2]. Equation (4) is consistent if dEs

is in nanovolts, Esa- in millivolts, and the "5's" are in parts per million. During a test, Va , V+, and V- are adjusted until the same EMF of the test TVC is obtained in each case when these voltages are applied

in sequence.

In the automated system, the test TVC's output EMF is held constant in the following manner. With a dc voltage applied

to the test TVC, an output EMF, £scl , is measured which is called the "set-point" voltage. In all subsequent applications of ac and dc voltages to the test circuit of Fig. 1 , the computer

is used to adjust the voltage applied so that the test TVC has an output very nearly equal to £sct . A four-step procedure is used for making ac/dc difference measurements. The proce- dure consists of applying a sequence of ac, +dc, —dc, and ac

voltages to the unit under test at approximately equal time

intervals. The following operations are performed for each step

in the procedure presently used with the system.

First, the nominal ac or dc voltage is applied, the test EMF,

£,, is measured, and a difference voltage, dE,, is calculated

as

dE, — £sci Et

. (5)

If the difference voltage is not sufficiently small, as discussed below, a voltage correction, dV,, is calculated from

Hence

dE,/n,Et = dV,/y,.

dV, = dE,V,/n,El

. (7)

The correction, dV,, is added to the nominal voltage, V„, and

a "reset" voltage, Vr = Vn + dV,, applied. The test EMF, £,,

is measured again, and a second voltage correction, dV2 , is calculated using (5) and (7). The output EMF of the standard TE, E„, is now measured. A correction for the EMF of the standard TE, dEs , is calculated, using dVj to compensate for any deviation of the actual EM F of the test TE from the set point £scl ; i.e., dEs = (ns Es dV2 )/Vr . (8)

The corrected EM F for the standard TE is then Esl =Es + dEs. (9)

b, =

Eg ~ E U ns Esd + 5, (10)

where Ea is the average of the standard's corrected EMFs

determined for the two applications of ac voltage; Esa-, «.t , and

bs are as defined in (4); and Ej is

Ed = Es \+ + £S | (11)

After completing the sequence of four steps by applying successively ac, +dc, —dc, and ac voltages, the ac/dc difference of the test TVC, 5,, is calculated as 544

where Es \+ is the corrected EMF calculated as in (9) when

+dc is applied, and £,,_ is the corrected EMF when —dc is applied.

In the computations previously discussed, account is taken

of the finite resolution of the ac and dc voltage sources. The

voltage setting remains unchanged if the EMF output of the

test TVC is close enough to the set-point voltage (£se t) that

a least significant digit change in the voltage source's output

will provide no decrease in the magnitude of the calculated

difference dE,.

A test run consists of four determinations of ac/dc differ- ence, and the average of the four determinations is calculated and printed on the system's printer. In general, three runs are normally made at each test voltage and frequency.

In order to increase the reliability of the EMF measure- ments made by the DV M, ten readings are automatically taken and the standard deviation of an observation of the ten readings

is calculated. If the deviation is >300 nV, the average EMF

value is discarded and the series of ten readings is repeated.

Up to ten attempts are made to obtain an average EMF value with a standard deviation <300 nV, and the test is aborted if the tenth attempt is unsuccessful. As the preceding equations show, the values of the exponent n for both the test and standard TVCs must be known. The

exponent varies with applied voltage, and its variation can be determined by doing "n tests." These tests will be discussed

in the next section. For the standard TEs, the coefficients in the equation expressing the value of n as a function of output

EMF are stored in a data file in the system software. Once the operator has specified which standard TE is being used, the standard TE's output EMF is measured with a +dc voltage

applied, and the value of n, is automatically computed. The

values of the test TE's coefficients for n must be manually

typed into the computer at the beginning of a test. To guard

against mistakes in entering data, the computed values of n for the test TE must be within the limits 1 .4 to 2. 1

An additional safeguard is contained in the ac/dc difference program to avoid application of voltages great enough to damage the TEs. When the reset voltage, Vr , is calculated, its value must be less than 1 20 percent of the rated voltage of the TVC. If the calculated reset voltage is outside this limit, the program aborts.

Other System Capabilities

As mentioned above, the system can readily be used to cal- ibrate ac voltage calibrators using TVCs with known ac/dc

differences. The output voltage of the test calibrator replaces

the output voltage of the system's ac source, and the test cali- brator's correction can be determined in terms of a calibrated
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT. VOL IM-32, NO 1, MARCH 1983

i

1 r

0 2 4 6 8 10 12 14 16 18 20 READING NUMBER

Fig. 2. DC voltage source stability, 10 V.

6 8 10 12 14 READING NUMBER

Fig. 3. AC vokage source stability, 10 V —20 kHz.

dc source and the known ac/dc differences of the standard

voltage converters used. The correction to the test calibrator (in ppm) is

1.90

£a — Ed C, = + bs + C<ic ns t d (12)

where Ea

is the measured EMF of the TVC with ac voltage

applied, Ed is the average EMF with direct voltage of both

polarities applied, ns is as previously defined, 6S is the correction (in ppm) to the standard, and C<jC

is the correction (in ppm) to the dc voltage source's nominal output obtained from

calibration data. As in the case of ac/dc difference tests, the

correcti6n is in parts per million if Ea — Ej is in nanovolts and

Ed in millivolts. To control the variability in the test data, it is important to make sure that the voltage sources are quite stable. Thus a program was written which uses the system's DVM to measure

the stability of the voltage sources with time.

Fig. 2 illustrates typical stability test data for the dc source. At a nominal setting of 10 V, 20 voltage readings were taken

over a time interval of about 20 min. The deviation of the measured voltage (in ppm) from the nominal voltage is defined

as AV

V

V, - V, X 106 (13)

where V\ is the measured value, and V2

is the nominal value. The maximum deviation measured was 1 .3 ppm and the mean

deviation was 0.6 ppm. The standard deviation of an observation was 0.5 ppm, and the 3cr limits for the mean were ±0.3 ppm. These results indicate that the dc source has acceptable

stability and is well within its specified accuracy of 50 ppm.

Typical results for a test of the ac source's stability are shown in Fig. 3. Again, 20 readings were obtained during a 20-min time interval. The test voltage was 10 V at a frequency

of 20 kHz. Here the range of deviations was 40 ppm with the mean being 1 9 ppm. The standard deviation of an observation was 9 ppm, and the 3er limits for the mean were ±6 ppm. The

discrete 10-ppm steps in the plotted data are due to the fact that the DVM's resolution on its 10-V range is limited to 10 ppm.

1.72

4 5 6 7 TE EMF IN mV

Fig. 4. Typical /j-test data.

As mentioned in the previous section, one important test that

is readily performed automatically is that of determining a TVC's characteristic. Typical results for a 2.5-mA,

VHF-type TE which was tested with a nominal 3-V range

resistor are illustrated in Fig. 4. The characteristic n is defined by

AE/E

AVIV

(14)

where AE is the measured change in output EMF for small changes in applied voltage AV, V is the nominal test voltage

for the TVC, and E is the measured EM F at the nominal test voltage. AV is programmed to be ±0.5 percent of nominal test voltage V. Sufficient time is allowed after each voltage change

for the TE to reach its final EMF value. For the results shown in Fig. 4, the TVC was tested from 50

to 1 10 percent of rated voltage. The total elapsed time to obtain the seven data points was about 1 .25 h. Each value of n is computed as the average of four deter- minations at any given voltage. A least squares analysis of the data provides the option of obtaining a fitted curve using up

to a fourth-degree polynomial. An analysis of variance table and residual standard deviation are computed and printed so

that the "goodness of fit" of the computed equation for n as

a function of TE output EMF can be determined. The printed

545
LENTER AND FLACH : AUTOMATIC SYSTEM FOR AC/DC CALIBRATION

750 I

1

1 1

1 ' 1' 1 1 r

PERCENT OF HATED V SECONDS

Fig. 5. Constant input reversal difference, A. Fig. 6. Typical response time of a TE.

data sheet also shows the values of ^(measured) — /j(computed). The figure shows an example of a linear fit which, in most cases, provides sufficent accuracy. The coefficients 2.00 and —0.04 are either stored in the computer's data file (for a standard TE) or are typed by the operator into the computer

program for the TE under test at the beginning of an ac/dc

difference test. The dc reversal difference (DCRD) of a TE can also be

readily determined. DCRD is defined as the difference in applied values of both polarities of dc voltage required to produce an equal output EMF (or "constant output") of the test TE

[7]. In order to facilitate DCRD measurements, a "constant

input" EMF, A, is actually measured. This A is related to the DCRD by a factor of 1 jn as

DCRD = (1/«)(A) =

(!/")

E+ - E. E+ + E-

(15)

where E+ and £_ are the TE output EMFs with equal +dc

and -dc (or "constant input") voltages applied, and n is as previously defined. During acceptance tests of some TEs, the data showed much

greater variation of DCRD from unit to unit than had been

expected. The only information available indicated that the DCRD should reach a maximum value between 70 to 80 per- cent of rated heater current and should approach a zero value

close to 100 percent of rated current. The results obtained did not substantiate this information. Hence, a program was

written to determine the DCRD as a function of applied volt- age. Typical results for a 5-mA, UHF-type TE tested with a 60-V range resistor are shown in Fig. 5. In this test, the voltage was varied from 50 to 1 20 percent of its rated value in 2-per- cent increments. For the sake of clarity in the figure, not all of the actual 36 data points are shown. It should be noted that the plotted data show A; hence, the DCRD values referred to a constant output EMF would be smaller by the factor of n (see

( 1 5)). In this test, the maximum A was about 700 ppm at 58

percent of rated voltage, and the minimum was about —330

ppm at 1 20 percent of rated voltage. At the time the DCRD testing program was developed, the

information on DCRD variation was limited. However, in the intervening time Inglis [8] has published a substantial amount

of information in this area. Investigations at NBS have not been as extensive as those of Inglis, but, in general, the results obtained are in agreement with his published data. The response time of a TE is also a characteristic which can

readily be measured using the automatic system. Response

time is obtained by making measurements of a TE's EMF

when full rated voltage is suddenly applied to the TE. The

calculated response time / is the time taken for the output

EMF to reach 1 — ( 1 /e) of its final value. Typical results for a 2. 5-mA, VHF-type TE are shown in Fig. 6. Measurement Results and Uncertainties

Manual calibration systems have been in operation for many

years at NBS, and their uncertainties have been documented

[l]-[3]. DC reversal difference, self-heating and ambient

temperature effects, drift effects due to nonequal time intervals between the EMF measurements, unbalanced lead impedances, induced voltages from electromagnetic fields, and operator fatigue and lack of skill are some of the more common

sources of error. The process of determining an estimated total uncertainty for a system is based upon 1) a history of measurement results; 2) careful measurements of standards with

accurately known corrections; 3) theoretical analysis of sources

of errors, along with, preferably; 4) intercomparison tests using two independent test methods [3]. A combination of 2) and

4) was used for the initial evaluation of the automated system.

The primary emphasis was placed on comparison of results obtained with a well-characterized manual system and those obtained with the new system.

A single-range (3-V), coaxial-type TVC was tested using

established manual techniques, followed by tests on the au- tomated system. The results obtained from the two test systems

are shown in Table I. For the automatic system, the values reported are the av- erage of 1 2 determinations of the transfer standard's ac/dc

difference. The random uncertainty was calculated as three times the standard deviation of the mean of the 12 observations, with the result that 3a = 21 ppm for the worst case. In each case, the differences between the results obtained with

546
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-32, NO. 1, MARCH 1983

TABLE I 3-V TVC AC/DC Difference (ppm)

Frequency

kHz Manual Automatic Di f ference

0.02 0 2 -2

20 1 -9 10

50 -1 -9 8

100 -1 -4 3

TABLE II 10-V TVC AC/DC Difference (ppm;1

Frequency

kHz Manual Automatic Ui f ference

20 -4 -17 13

50 -3 -7 4

100 -1 6 -7

the two test methods are less than the total uncertainty of the manual test system, which is 20 ppm for all voltages for frequencies in the range 0.02-20 kHz, 30 ppm for 20-50 kHz,

and 50 ppm for 50-100 kHz (the lower uncertainty applies at the crossover frequencies).

A second test was made using a 10-V range TVC as a

transfer standard. The test results are listed in Table II. Both the 3-V and 10-V TVC transfer standards had small ac/dc differences. Therefore, a third intercomparison test was

conducted using a transfer standard with large, but accurately

known differences. The results of this test are listed in Table

III.The test results summarized in the three tables suggest a possible systematic uncertainty since the automatic system

gave lower values for all but three test frequencies. Further

investigation is required to evaluate whether or not this uncertainty is significant, and a multirange transfer standard is presently being evaluated.

Conclusions

The results indicate that it is feasible to make fully auto- matic precision ac/dc difference measurements at one voltage. The principal advantage of the system is the elimination of the need for any type of manually or automatically balanced TE

comparator, permitting the system to be assembled using

commercially available instrumentation, except for the switching modules. This approach, therefore, should make

automation attractive to many calibration laboratories faced

with the need to make ac/dc difference measurements.

Moreover, the system can be used as a nearly self-calibrating

TABLE III 50-V TVC AC/DC Difference (ppm)

Frequency

kHz Manual Automatic Di f ference

5 29 26 3

10 56 52 4

20 121 121 0

50 320 310 10

meter calibration system. Measurements can be done with minimal operator intervention, eliminating tedious and fa- tiguing manual operations. Hence, personnel presently en- gaged almost exclusively in manual ac/dc difference tests can

be freed to do other work. Experience obtained during many

years of testing with manually operated systems at NBS has shown that the precision of the test results is dependent upon

operator skill. The automatic system's precision is independent

of operator skill, eliminating this variable in the experimental

uncertainty.

ACKNOWLEDGMENT

The authors are indebted to M. S. Carter who wrote much

of the software and solved many difficult programming

problems encountered in developing the system. S. G. Tre- maine, L. Freeman, and E. V. Byland also made valuable

contributions to some of the programs which have become part of the total system software. C. B. Childers helped greatly by

doing the measurements on the manual system. Acknowledgment goes to B. A. Bell for his conceptual ideas and continuing support in the development of this system.

References

[1] F. L. Hermach and E. S. Williams, "Thermal converters for audio-fre- quency voltage measurements of high accuracy," IEEE Trans. Instrum

Meas., vol IM-15, pp. 260-268, Dec. 1966.

[2] E. S. Williams, "Thermal voltage converters and comparator for very accurate ac voltage measurements," J. Res. Nat. Bur. Stand. (U.S.), vol. 75C, pp. 145-154, Dec. 1971.

[3] F. L. Hermach, "AC-DC comparators for audio-frequency current and

voltage measurements of high accuracy," IEEE Trans. Instrum. Meas.,

vol. IM-25, pp. 489-494, Dec. 1976.

[4] R. S. Turgel, "A comparator for thermal ac-dc transfer standards," ISA

Trans., vol. 6, no. 4, pp. 286-292, 1967.

[5] K. J. Lentner el al., "A semiautomatic system for ac/dc difference cali- bration," IEEE Trans. Instrum. Meas., vol. IM-29, pp. 400-405, Dec.

1980.

[6] E. S. Williams, "A thermoelement comparator for automatic ac-dc dif- ference measurements," IEEE Trans. Instrum. Meas., vol. IM-29, pp. 405-409, Dec. 1980.

[7] "American national standard for ac-dc transfer instruments and converters," ANSI C100.4— 1973, Amer. Nat. Stand. Inst., Inc., 1430 Broadway, New York, NY 10018.

[8] B. D. Inglis, "Errors in ac-dc transfer arising from a dc reversal differ- ence," Metrologia, vol. 1 7, pp. 111-117, 1981.

547
AC VOLTAGE CALIBRATIONS FOR THE 0.1 Hz TO 10 Hz

FREQUENCY RANGE

Howard K. Schoenwetter

The development of voltmeters to meet the need for rms voltage measurements in the infrasonic frequency range is

discussed as well as the need to trace these measurements to

the U.S. legal unit of voltage. A new method for supporting

voltage measurements in the 0.1 Hz - 10 Hz range was described

in a 1979 paper and is discussed further. The principles of

the method are embodied in detailed procedures given for

calibrating sine-wave voltage standards and rms voltmeters

over the 0.1 Hz - 10 Hz frequency range, using the NBS AC

Voltmeter/Calibrator. The sine-wave calibrator of this

instrument, used for these calibrations, has an accuracy of

0.020 percent over the 0.5 mV - 7 V range.

Key words: ac voltage calibrations; ac voltage calibrators;

ac voltage standards; infrasonic voltage measurements;

low-frequency voltage measurements; rms voltmeters.

1. INTRODUCTION

Calibrated vibration transducers are required in laboratories

and test facilities to determine the acceleration of vibration exciters,

tables, and fixtures. They are also used to measure the vibrations

in many other types of machines and structures, including prototype

models of military, space, and commercial vehicles undergoing qualification

testing. In past years, calibrations of these transducers at frequencies

below 2 Hz were either omitted or performed with uncertain accuracy.

This was because the voltmeters available for measuring the transducer

outputs at these frequencies had questionable accuracy and excessively

long response times (15 to 20 periods of the measured voltage).

In 1976, an rms digital voltmeter (DVM) was developed at the

National Bureau of Standards (NBS) to support vibration measurements

over the 0.1 Hz - 50 Hz range J It was designed to measure voltages

from 2 mV to 10 V with an uncertainty of approximately 0.1 percent

of reading. The maximum response time of 40 seconds corresponds to

four periods of the lowest frequency signal that can be measured.

Since the means for calibrating the voltmeter to the required accuracy

did not exist, a voltage cal i brator was developed and incorporated

iThis work was supported, in part, by the DC and Low Frequency

Calibration Coordination Working Group of the Department of Defense.

548
into the same instrument. This calibrator, providing bipolar dc voltages

and sine-wave voltages at frequencies of 0.1, 0.2, 0.5, 1, 2, 10, and

50 Hz, was intended not only for calibration of the companion voltmeter

but also for the calibration of other low-frequency voltmeters.

During the 1976-77 period, the NBS AC Voltmeter/Calibrator was

used at the Metrology and Calibration Centers of Redstone Arsenal (Army)

and Newark Air Force Base, as well as the Vibration Section at NBS,

to measure vibration transducer outputs and to calibrate other ac voltmeters [1]^. Except for the need of a 5-Hz calibration frequency,

the instrument met all present and expected future requirements for

support of voltage measurements at very low frequencies. ^ Consequently,

a detailed instruction manual and an archival paper were written to

facilitate reproduction of the instrument [2]. A commercial instrument

based on the NBS design later became unfeasible because the ac voltmeter

portion utilizes a multi -juncti on thermal converter (MJTC), which

has been available only in limited quantities from one manufacturer.

Subsequent to the development of the NBS AC Voltmeter/Calibrator, at

least two other instruments have been developed which can measure

voltages at infrasonic frequencies. NBS has developed a Low Frequency

AC Samping Voltmeter which has approximately 0.1 percent uncertainty

and a nominal maximum response time of two periods of the lowest

frequency signal (0.1 Hz) that can be measured [3,4], 5 Also, a large instrumentation manufacturer has developed a multimeter with a sampling voltmeter, operable down to 0.1 Hz or lower. However, the

accuracy and response time of the voltmeter depend upon the frequency

of the measured voltage.

As the number of ac voltage measurements in the 0.1 Hz - 10 Hz

range increases, because of the greater availability of suitable

voltmeters, the need increases for tracing these measurements to the

U.S. legal unit of voltage. Selecting the best method for effecting

this traceability is discussed next.

^Numbers in brackets refer to the literature references listed at

the end of this report.

^This calibration frequency was added later.

^Commercial voltage calibrators operate at frequencies down to 10 Hz

and can be calibrated using conventional ac-dc transfer measurements.

Also, a number of commercial voltmeters function satisfactorily above

10 Hz; therefore, the voltage measurement and calibrator capability

above 10 Hz were not considered vital.

^Six of these units have been made by NBS, and supplied to each of

the DoD metrology laboratories. They were calibrated by using the

NBS AC Voltmeter/Calibrator as a reference standard.

549
2. A METHOD FOR INFRASONIC VOLTAGE MEASUREMENTS

Standards laboratories calibrate ac voltage standards against

dc voltage standards, using thermal voltage converters (TVCs) to make

the ac-dc voltage comparisons. Calibrations of the dc voltage

standards and TVCs are traceable to reference standards maintained

by NBS. Theoretical and practical limitations arise in calibrating

TVCs (measuring their ac-dc difference), and employing them to make

calibrations at the lower infrasonic frequencies [5]." These

limitations cause the calibrations to be slow, costly, and inaccurate,

particularly at frequencies below ~2 Hz. For these reasons, a new method was proposed in [5] for supporting ac voltage measurements

below 10 Hz, which supplements the existing method as shown in figure 1.

The primary objective of the new procedure is to make it feasible to

equip standards laboratories with easily calibrated multi -range sine-wave

standards, which operate at frequencies of 0.1, 0.2, 0.5, 1, 2, 5,

and 10 Hz. These voltage sources would be used mostly to calibrate

voltmeters at these frequencies over the voltage range of approximately

0.5 mV - 7 V rms.7

The proposed equipment and procedures for testing a multi -range

voltage source are described in sections IV and V of [5]. Central

to the method is a single voltage level sine-wave reference source with frequencies of 0.1, 0.2, 0.5, 1, 2, 5, and 10 Hz, whose amplitude

variation with frequency (frequency response) is calibrated and has

excellent long-term stability. This unit serves as a voltage transfer

standard. By adjusting the output of the multi -range source to be

approximately the same as the voltage level of the reference, the

frequency response of the source can be determined from ac-ac comparisons, using an rms transfer voltmeter. Following this step,

the voltage level at each frequency is established by calibrating

the unit at 10 Hz, using a dc voltage standard and a TVC to make an ac-dc transfer measurement.

Except for the rms voltmeter contained in the NBS AC

Voltmeter/Calibrator, existing rms voltmeters do not have the necessary

high resolution and small differential nonlinearity to make ac-ac comparisons for calibration work. ^ Until rms voltmeters suitable

6 For convenience, reference 5 is included at the end of this report.

^Present commercial ac voltage calibrators either have insufficient

accuracy over this voltage range or only operate above 1U Hz.

fy\ transfer voltmeter must be able to measure voltage differences as small as 20 ppm of the measured value, both voltages being of the same frequency. The ac-dc difference and accuracy of the voltmeter

are not critical, however.

550
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551
for this work become available, calibration of multi -range sine-wave

voltage sources 9 and rms voltmeters will be performed at NBS.

Procedures for these calibrations are discussed in sections 4 and 5.

Since the ac calibrator portion of the NBS AC Voltmeter/ Cal i brator

(see fig. 1) would be a suitable multi-range voltage standard for use

by standards laboratories, the detailed design of this circuit is

given in appendix B. The methods used to calibrate the unit are included in the calibration procedure described in the next section.

3. CALIBRATION OF NBS AC VOLTMETER/CALIBRATOR

As described in [2] and [5], the NBS AC Voltmeter/Calibrator

consists of an rms digital voltmeter and a calibrated voltage source which can function as either a dc or ac voltage calibrator. The rms voltmeter consists of an input amplifier, rms/dc converter, and dc UVM.

The ac voltage calibrator is used to calibrate ac voltmeters and

is used in conjunction with the rms voltmeter to calibrate ac voltage

sources. The procedure used to calibrate the ac voltage calibrator

is shown in figure 4 of [5], To perform this calibration, the

frequency response of the ac calibrator and the ac-dc difference

of the rms/dc converter must be known from prior measurements. All

of these calibrations are listed below and are discussed in this

section

(1) Frequency response and voltage calibrations of the ac voltage

cal i brator.

(2) Calibration of the laboratory dc voltage standard (standard cell).

(3) Calibration of the dc calibrator.

(4) AC-DC difference calibration of the rms/dc converter.

The errors associated with the items listed above all contribute to

the estimated uncertainty assigned to the ac calibrator voltage and

will be summarized after these calibrations have been described.

Several refinements were made recently to simplify the task of

calibrating the AC Voltmeter/Calibrator and to improve its overall

performance. These changes are also discussed in this section.

^Assuming that the voltage standards employ a D/A converter for the

sine-wave generation, a frequency response (relative voltage) calibration

is preferable to a conventional voltage calibration.

552
3.1 Calibration of AC Voltage Calibrator

Calibration of the ac voltage calibrator consists of two

calibrations: (1) a frequency response calibration in which the

calibrator voltages at 5, 2, 1, 0.5, 0.2, and 0.1 Hz are compared

with the voltage at 10 Hz, and (2) a voltage calibration at 10 Hz.

3.1.1 Frequency Response Calibration of AC Calibrator

The frequency response of the ac calibrator was originally

determined using peak-to-peak and rms measurements, the latter being

made with the rms voltmeter and limited to the 1 Hz - 10 Hz range [2].

Average-value measurements were later combined with the above results [5].

Since the peak-to-peak method uses a storage oscilloscope to exhibit

the voltage peaks, the measurements are somewhat subjective and are considered less reliable than the average-value measurements; therefore,

only the latter (along with limited rms measurements) are now used

for the frequency response calibrations. The precision rectifier-filter

circuit used for these measurements is represented by figure 5 of [5],

This circuit and associated power supply are contained in a separate

shielded enclosure. The justification for basing the rms frequency

response of the ac voltage calibrator on average-value (or peak-to-peak)

measurements is given in [5].

Figure 2 shows the front panel operating controls of the NBS AC

Voltmeter/Calibrator. The 5 Hz and 0.1 Hz frequencies are selected

with the CAL FREQ switch in the 10 Hz and 0.2 Hz positions, respectively,

using the rear panel "5 Hz-10 Hz" and "0.1 Hz-0.2 Hz" switches (see fig. 3).

For average-value measurements, the precision rectifier circuit is

connected between the CAL 00T terminals and the AVE. VOLT. IN terminals

(rear panel) with the PREC RE CT-INTERNAL switch in the PREC RECT

position. In this switch position, the rms/dc converter is disconnected

and the output of the precision rectifier is connected to the input

(point o) of the voltmeter's low pass filter (see fig. 2 of [5]).

A dc voltage level of 5 V (nominal) at this terminal corresponds

to a full-scale indication of the voltmeter. Therefore, the full-scale

voltmeter reading is normalized to whatever range is selected by the

V-RANGE-MV switch. Since full-scale readings of 10000.0, 5000.0,

2000.0, 1000.00, —- result from range selections of 10, 5, 2,

1, — volts, it is seen that the best resolution is obtained with

the 10, 1, 0.1 or 0.01 V ranges. (Unless indicated otherwise, the

PERIOD is set to 10 for all measurements.) The 10 V range has been

arbitrarily selected for the frequency response measurements. Because

the rms/average ratio for a sinewave is ~1 . 1 1

, the ac calibrator

should be set to approximately 5.55 V to yield a full-scale output

of 5 V from the rectifier-filter circuit. (The "fl00" attenuator

switch, not shown in fig. 2, should be kept in the OUT position.

Also, the FUNCTION switch should be in the MEAS position.)

553
554
HEAVY FILTERING OFFc@ ON

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CAL FREQ RESPONSE OVM FREQ RESPONSE

0.1 0.2 0.5 1 2 5 (0) cS) (6)

LEFT SIDE PANEL

gure 3. Controls and connectors on rear, top, and si

panels of AC Voltmeter/Calibrator.

555
The dc DVM employed in the rms voltmeter of the NBS AC

Voltmeter/Calibrator (fig. 2 of [5]) consists of a V/F converter

and a counter-timer which counts the output pulses from the V/F

converter over periods of 1 s or 10 s (PERIOD set to 1 or 10),

determined by a crystal -control led oscillator. An integrating

voltmeter was used to effectively reduce the ripple in the output

voltage from the low-pass filter, to minimize fluctuations in the

displayed voltage. The frequency of the ripple voltage in the outputs

of both the rms/dc converter and the precision rectifier circuit is

twice the frequency of the input voltage to these circuits. If the

counter period (integration period) includes an integral number of

ripple periods, the ripple in the dc voltage applied to the dc DVM

will integrate to zero [2]J0 Therefore, fluctuations in the indicated

voltage from ripple are zero for frequencies of 0.05, 0.1, 0.15, — Hz,

which include the nominal values of the calibrator frequencies. (An

integration period of 10 s must be used for frequencies of 0.5 Hz

and less.) To allow for some deviation from these frequencies by

the AC Voltage Calibrator, ripple filtering in the low-pass filter

and rms/dc converter can be increased (with respect to the original

design) using the HEAVY FILTERING switch on the rear panel of the

AC Voltmeter/Calibrator.

The effect of this filtering on rms measurements will be discussed

later. When the precision rectifier is used, the additional filtering

limits the measured peak-to-peak ripple voltage to ±0.30 percent of

the dc voltage applied to the dc DVM when the input frequency is near

0.1 Hz. (The ripple voltage is much less near the higher calibrator

frequencies.) The computed maximum peak-to-peak fluctuation in the DVM

readings is approximately ±30 ppm for each percent deviation in frequency

from 0.1 Hz. For all calibration work, the calibrator frequencies should

be held within ±0.05 percent of their nominal values so that the effect

of ripple on DVM readings is negligible (±1.5 ppm or less, based on the

preceding statements). The calibrator frequencies are at their nominal

values when the clock frequency, from which they are derived, is 6400 Hz.

The clock frequency is easily held with ±0.05 percent of 6400 Hz, since

observed daily frequency variations are usually less than ±2 Hz (±0.03%).

The frequency can be adjusted by the CLK ADJ pot and is displayed by the

counter-timer when the COONTER switch is in the CLK position (see fig. 3).

Worst-case DVM reading errors caused by ripple voltage can be

avoided by using the average of several successive readings instead

of a single reading, since the ripple integrations in the dc DVM will

vary in size and usually are of both polarities. Therefore, the

average value is more accurate than the worst-case single reading

l^in other words, the ripple voltage integrates to zero if a time

interval equal to two counter periods includes an integral number of

input signal periods.

556
value. Unless indicated otherwise, maximum filtering (FILTER FREQ

switch in position L and the HEAVY FILTERING switch on) should be

used for all calibration work. With maximum filtering, the DVM

settling time to within 5 ppm is about 2 minutes. This response time

must be -considered when devising the measurement procedure for

determining the calibrator frequency response.

Table 1 shows a typical set of average-value measurements which

was used to evaluate the frequency response of the ac calibrator.

The DVM readings are represented by Vf or V]q, where the subscripts

denote the frequency of the voltage being measured. Calibrator

frequencies are switched approximately every 3 minutes, allowing

2 minutes for voltmeter settling and 1 minute for recording six readings.

If the first-recorded value is not repeated, it should be discarded,

since this probably indicates lack of complete settling by the DVM.

Successive numbers represent the least significant digit of successive

DVM readings. The 0.5, 0.2, and 0.1 Hz measurements are compared with

the succeeding 10 Hz measurement to minimize the time periods between

readings that are compared, thus minimizing the effect of voltmeter drift.

Table 1. Typical set of average-value measurements used to

evaluate the frequency response, Vf - V-jq, of the

ac voltage calibrator. The nominal value of

Vf and Viq is normalized to 10U0 units.

Elapsed Mean

time(min) Frequency(f) DVM reading(V f

) a value V

f -V

0 10 Hz 1000.47,8,8,8,8,7 1000.477

3 5 1000.48,8,8,8,8,8 1000.480 +0.003

6 2 1000.48,8,8,8,8,8 1000.480 +0.003

9 1 1000.48,8,8,7,8,8 1000.478 +0.001

12 0.5 1000.48,8,9,8,9,8 1000.483 +0.005

15 0.2 1000.48,8,8,9,8,8 1000.482 +0.004

18 0.1 1000.48,7,7,7,8,8 1000.475 -0.003

21 10 1000.48,8,7,8,8,8 1000.478

Successive numbers represent the least significant digit of successive

readi ngs

.

557
Erratum: Third line, third paragraph should be 12 degrees of freedom

instead of 2 degrees

For routine cal ibration s of the frequency response, it is planned

to use the averages (Vf - V-jq) from two sets of measurements like those

shown in table 1. In order to hold the magnitude of (Vf - V-|q) near

z ero, freq uency response adjustments have been added recently. If

(Vf - V

-j q) exceeds ±10 ppm for any frequency, the appropriate CAL

FREQ RESPONSE pot should be adjusted to minimize the voltage difference

(see fig. 3)

The estimated standard deviation of the measurement process is based

on 15 sets of measurements, divided evenly among three groups. One group

of measurements was made after the frequency response adjustments were

added. A second group was made before the adjustments were provided.

The third group of sets was obtained from the Naval Air Test Center,

Patuxent River, MD (NATC) calibrator, which is based on the NBS

design without frequency response adjustments. The maximum magnitude

of (Vf - V]q) was approximately 5, 27, and 12 ppm, respectively, for

the first, second, and third groups of sets. Although the statistical

populations corresponding to the three groups have different values

of (Vf - V]o), their standard deviations should be approximately

equal, since all three calibrators had nearly identical electrical

and mechanical designs. Therefore, a worst-case pooled estimate of

the standard deviation (Sp), based on these three groups (15 sets)

of measurements, will be used as the estimated standard deviation of

a (Vf - V

-j q

) measurement. The largest pooled estimate of the

standard deviation is 4.4 ppm and occurs for a frequency equal to

0.5 Hz.

The estimated imprecision in (Vf - V -j o )

» obtained from N sets of

measurements, can be expressed by (s

p //N)t, where t is Student's t,

based on 2 degrees of freedom and a selected confidence level of

0.99. If (Vf - V-|~q~) is determined from two sets of measurements, the

calculated imprecision is approximately 10 ppm. As mentioned

previously, Vf will be readjusted if (Vf - V \q) is larger than

±10 ppm. A control chart (f.i_g._ 4) will be kept showing calibration

dates and the value of (Vf - V^q] before being reset.

3.1.2 Voltage Calibration of AC Calibrator

The voltage level of the ac voltage calibrator is established by

comparing the ac calibrator voltage at 10 Hz with the dc calibrator

voltage, using the rms voltmeter as a transfer standard. For these

ac-dc comparisons, the PREC RE CT-INTERNA L switch is set to the

INTERNAL position, HEAVY FILTERING is switched to OFF, and the FILTER

FREQ switch set to L. Also, the PERIOD is set to 10 to yield maximum

voltmeter resolution, and the DVM FREQ RESPONSE switch should be in

the OUT position.

Before making ac-dc comparisons, the dc offset voltage of the

calibrator buffer amplifier should be adjusted to a minimum value as follows. Set the CAL VOLTS dial to all zeros, the RANGE switch to 1 and the FUNCTION switch to INT AC CAL. Then, using a sensitive null

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detector, adjust the XI OFFSET for a minimum voltage (<1 yV) at

the CAL OUT terminals. With the RANGE at 10, adjust the X10 OFFSET

for a minimum voltage (<10 yV). Following these adjustments, set

the CAL FREQ to 10 Hz, CAL VOLTS to 5.000000, and the V-RANGE-MV switch

to 10 V. Also, connect the CAL OUT terminals to the TE INPUT terminals

with a coaxial cable and set the TE switch to EXT. Compare the ac calibrator voltage with the dc calibrator voltage (average of both

polarities) as shown in table 2.

Table 2. A comparison between ac and dc calibrator voltages,

applied directly to the rms/dc converter (TE INPUT).

The FUNCTION and CAL columns are switch positions on the AC Voltmeter/Calibrator. The averages of ac and

dc mean values are used for V ac and V dc . The nominal

value of the DVM readings is normalized to 1000 units.

Elasped FUNCTION Mean

time(min) INT CAL CAL DVM readings value V ac

- V dc

AC 1006.64,4,4,4,3,4 1006.638

0.6365-0.6236

DC + 1006.64,4,4,4,3,3 1006.637 = 0.013

6 DC - 1006.61,1,1,1,1,1 1006.610

9 AC 1006.64,3,4,3,4,3 1006.635

For rou tine ac-dc difference calibrations, it is planned to use the average (

ar - V^T) from three comparisons like the one shown

in table 2. If (V a

<- - V dc ) exceeds ±20 ppm, the AC REF GAIN should

be adjusted for a minimum difference. The estimated standard deviation s of a comparison, is 7.0 ppm, obtained from 13 compa risons. Us ing the

expression (s//N")t to calculate the imprecision of (V ac

- V^)

yields approximately 12 ppm, where N is 3 and Student s t is based on

12 degrees of freedom and a confidence level of 0.99. Since (Vac

- V dc ) will be readjusted if it is larger than ±20 ppm, the maximum estimated

systematic error is ±20 ppm. A control chart (fig. 5) will be kept

showing calibration dates and the value of the ac-dc difference before

the AC REF GAIN is readjusted.

The dc offset of the function generator used in the sine-wave

reference should be checked approximately every three years. It can

be measured at pin 11 of the function generator with the DPDT switch

in the ZERO position (see fig. B-2). The switch and offset adjustment

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(R21 ) are accessed by removing the top panel. Adjustment of the offset

is non-critical and is seldom required. Measurements show that an offset

as large as ±200 y V contributes less than ±1 ppm to the rms value

of the 1 V ac reference. A dc offset has a much larger effect on the

average value of a sine wave; however, an offset voltage affects all

calibrator frequencies the same and has no affect on frequency response

measurements.

The measured temperature coefficient of the ac voltage reference

is approximately +3 ppm/°C. Therefore, the imprecision for a 2°C range

is ±6 ppm.

3.2 Calibration of the Standard Cells

The basic laboratory voltage standard to be used for calibrating the

dc voltage calibrator is a set of four saturated standard cells,

housed in a temperature-controlled standard cell enclosure and located

near the AC Voltmeter/Calibrator. The cell voltages are fed to the

NBS Volt Facility via a cable to facilitate in situ calibration. It

is planned to calibrate the cells at least once a year. Calibrations

performed in January and November of 1982 show a maximum drift of

0.4 ppm for this 10-month period.

3.3 Calibration of the DC Calibrator

The voltage of one of the calibrated standard cells (approx.

1.01815 V) is transferred to a "working" reference, using a null detector for the voltage comparison. A stable, 10 V laboratory

reference with seven decade resolution (1 yV) is suitable for the

working reference. The null detector should have approximately 140 dB common mode rejection and at least 1 \An input impedance. The

detector is used for all voltage comparisons and offset voltage (zero)

measurements described in this section.

A laboratory Kel vi n-Varley (K-V) divider is used to "step" the

reference voltage to the appropriate level for calibrating the dc

reference, buffer amplifier, and -=-100 attenuator of the calibrator.

The K-V divider should have seven-digit resolution and a linearity

of 0.1 ppm of input for dial settings between 1.0 and 0.1.

To calibrate the dc reference and buffer amplifier gain, set the

CAL VOLTS dial to 0000000 and the FUNCTION switch to INT AC CAL.

With the RANGE at 1, adjust the XI OFFSET for minimum voltage

(<1 P V) at the CAL OUT terminals. With the RANGE at 10, adjust

the X10 OFFSET for minimum CAL OUT voltage (<10 UV). Return the

RANGE to 1 and set the FUNCTION switch at INT DC CAL, CAL at + and

CAL VOLTS at .99999910 (last digit set to 10). Set the K-V divider

to the reciprocal of the standard cell voltage (approx. .9821700) and

connect its input to the laboratory reference. Then, adjust the DC

REF pot so that the calibrator voltage equals the K-V divider output

562
voltage. The gain of the buffer amplifier is adjusted with RANGE at

10, CAL VOLTS at 5.000000, and the K-V divider set to the ratio of the

standard cell voltage to 5 V (approx. .2036300). With the K-V divider

input connected to the CAL OUT terminals, adjust the X10 GAIN so that

the divider output voltage equals the laboratory reference voltage.

In order to minimize errors in the ac and dc calibrator voltages

at small signal levels (< 50 mV) caused by extraneous signal and

noise pickup, voltage offsets and Kel vin-Varley divider (see fig. 2 [5])

nonlinearity, an attenuator has been provided which attenuates

both ranges of the dialed voltage by a factor of 100. This precision

resistive attenuator has a 50 n output impedance and is inserted

between the buffer amplifier output and the CAL OUT terminals when

the attenuator switch is in the "*100" position. When calibrator

voltages larger than 50 mV are desired, this switch should be in the

OUT position. If required, calibration of the VI 00 attenuator

should immediately follow the preceding calibrations. The calibrator

control positions are not changed, except for putting the attenuator

in the *100 position. The laboratory K-V divider input is connected

to the laboratory reference and set to the ratio of 0.05 V to the

standard cell voltage (approx. .0491090). The *100 pot is then

adjusted so that the calibrator voltage equals the K-V divider output

voltage.

The uncertainty in transferring the standard cell voltage to the

laboratory voltage reference at the 1.01815 V level is less than

±2 ppm. The uncertainties in stepping the reference voltage

down to 1.000000 V (calibration of dc reference) and up to 5.000000 V are 4 and 10 ppm, respectively. These uncertainties are caused

mostly by varying thermal voltages at connectors, circuit noise, and

lack of resolution in adjustments. The 10 ppm uncertainty includes

1 ppm systematic error, caused by the nonlinearity of the laboratory

K-V divider. The measured temperature coefficients of the dc reference

and buffer amplifier (on X10 range) are -6 ppm/°C and +2 ppm/°C,

respectively. The corresponding imprecisions for a 2°C range are

±12 ppm and ±4 ppm.

The expected maximum temperature coefficient for the attenuator is

±3 ppm/°C (±6 ppm uncertainty for a 2°C range). Tests show

that thermal voltages up to ±0.7 y V are generated at the CAL OUT

terminals when the voltage applied to the t100 attenuator is zero.

This corresponds to ±14 ppm of the 50 mV output from the attenuator

when it is being calibrated. Also, the nonlinearity of the laboratory

K-V divider at the .0491090 setting may contribute 2 or 3 ppm error to

the calibration of the attenuator. Finally, the adjustment of the

attenuator has an uncertainty of about 2 ppm. Therefore, a systematic

error of 3 ppm (from K-V divider) and an imprecision of 22 ppm have

been assigned to the vl00 attenuator calibration.

563
Based on a linearity comparison with the laboratory K-V divider,

the K-V divider employed in the calibrator can cause up to 25 ppm error in the ac calibrator voltage (V

c

). It is planned to check this

unit against the laboratory standard approximately once a year.

Control charts (fig. 5) will be kept which show the changes in

the ac and dc reference voltages and the buffer amplifier gain that occur between calibrations. Based on measurements and manufacturer's

specifications, the yearly drift rates of the standard cell, 4-100 attenuator, and the K-V divider are expected to be less than 2, 2,

and 3 ppm, respectively. Since these systematic errors are very small

compared with the expected uncertainty of the ac calibrator voltage,

no control charts are planned for these components.

3.4 AC-DC Difference Calibration of RMS/DC Converter

The ac-dc difference of the rms/dc converter used in the rms voltmeter was calibrated at NBS in April 1981. The reference thermal

voltage converter (TVC), employed in the calibration test set,

consisted of model FE reference thermal element and model F.10V (#1)

multiplier range resistor. This thermal element, consisting of four

single junction thermal elements in series, was designed to have

negligible ac-dc difference for frequencies down to 10 Hz or below.

Ten comparisons were made between the rms/dc converter and reference

TVC involving four measurements each (ac, dc, dc reverse, ac), yielding

an average ac-dc difference of ±0.25 ppm. The standard deviation

of a comparison was 5.83 ppm, so that the standard error of the mean

was 1.84 ppm. If the selected confidence level is 0.99, Student's t is 3.3 and the imprecision in the calculated ac-dc difference is

±6 ppm.

The only known source of systematic error in the ac-dc calibration

is the uncertainty in the ac-dc difference of the reference TVC.

This unit has been i ntercompared with several other reference TVCs,

with an ac-dc difference of less than 1 ppm [6], These units are maintained at NBS and are believed to have zero ac-dc difference to

within 1 or 2 ppm for frequencies as low as 10 Hz. If ±2 ,ppm

uncertainty is allowed for the reference TVC, the estimated ac-dc

difference of the rms/dc converter is 0 ±8 ppm.

It is recommended that multi -juncti on thermal converters (MJTCs)

be re-calibrated every five years. This re-calibration interval should

also apply to the rms/dc converter, since its ac-dc difference is

determined by the ac-dc difference of the MJTC employed.

3.5 Summary of Calibration Errors of AC Voltage Calibrator

Table 3 lists the estimated uncertainties which limit the

calibration accuracy of the ac calibrator. Item 9 of this

table 3 shows that, following a frequency response calibration, the

(Vf-V-|Q = 0) with an uncertainty of ±20 ppm. It should be emphasized

564
Table 3. . Estimated systematic error and imprecision components

of the uncertainties which limit calibration accuracy

of ac voltage calibrator. Estimated components of

uncertainties less than 0.5 ppm are rounded to zero.

Systematic

error Impreci si on

(1) Standard cell uncertainty 2 ppm U ppm

(2) Standard cell lab reference transfer 0 2

(3) Step-down to 1 V level in dc reference 0 4

(4) Step-up to 5 V level in dc calibrator 1 9

(5) Temperature coefficient of dc reference

(-6 ppm/°C) for a 2°C range 0 12

(6) AC/DC comparisons (calibration of ac calibrator at 10 Hz) 20 12

(7) AC-DC difference uncertainty of

rms/dc converter 2 6

(8) Temperature coefficient of ac reference

(+3 ppm/°C) for a 2°C range 0 6

(9) Calibration of (Vf-Vio) 10 10

(10) Maximum error from calibrator K-V divider 25 0

(11) Maximum K-V drift between calibrations 3 0

(12) Temperature coefficient of buffer

amplifier on XI 0 range

(+2 ppm/°C) for a 2°C range 0 4

(13) Calibration of the *100 attenuator 3 22

(14) Maximum attenuator drift between

calibrations 2 0

Sub-totals 68 33 (rss)

Total uncertainty of ac calibrator voltage V c ±101 ppm

565
that when frequency response calibrations are made of a customer's

voltage source or voltmeter, only the uncertainty of ( -V

-j q

) needs

to be known. Table 3 also shows that the estimated uncertainty of

the ac calibrator voltage V

c is ±101 ppm, following a complete

calibration of the dc and ac calibrator circuits.

Although periodic calibration intervals cannot be established for

these circuits until control chart data are available, it is believed

that a specified maximum uncertainty of ±200 ppm for the ac calibrator

will allow calibration intervals approaching one year for the dc

reference voltage, buffer amplifier gain, and ( Vf -V -jq) . A somewhat

shorter calibration interval is expected for the ac reference. The

corresponding error budget for (Vf-V-|o) alone is ±35 ppm. Therefore,

the calibration intervals will be chosen to ensure that the maximum

uncertainties of ( V^-V -j q

) and V

c are less than ±35 ppm and

±200 ppm (0.020%), respectively.

Although considerable reliance will be placed on control chart

data to anticipate future drifts in calibrator circuits, accurate

checks on circuit performance can be made at any time using shortened

procedures. Using only one set of measurements as shown in table 1 to determine (Vf-V]g), instead of the average of two sets, decreases

the measurement time from 48 minutes to 24 minutes. The imprecision

increases from ±10 ppm to ±14 ppm and the allowable value of

( Vf-V ] o ) should be increased from ±10 ppm to ±14 ppm before being

readjusted. Using one set of measurements as shown in table 2 to determine

(V ac -V (j C

), instead of the average of three sets, decreases the measurement

time from 36 minutes to 12 minutes. The imprecision increases from

±12 ppm to ±21 ppm and the allowable value of (V ac

-V ^c

) should

be increased from ±20 ppm to ±35 ppm before being readjusted.

Substituting these values into table 3 (items 6 and 9) yields maximum

uncertainties of ±28 ppm and ±125 ppm for (V^-V-jq) and V c , respectively. The time required to calibrate the dc reference, buffer

amplifier gain, and 4-1 00 attenuator is 13 minutes. (These procedures

cannot be shortened.) Therefore, the total calibration time of the

dc and ac calibrators, using the shortened procedures, is approximately

50 minutes.

3.6 Frequency Response of the RMS Voltmeter

The ac calibrator can be used to calibrate voltage standards at

frequencies other than the nominal calibrator frequencies, if the

frequency response of the rms voltmeter (DVM) is known. The response

of the DVM decreases as the frequency of the input signal is decreased

below ~2 Hz, because of increasing ripple voltage in the rms/dc

converter and lack of square law response by the MJTC of this circuit [2j.

The effect from ripple is minimized, however, by the HEAVY FILTERING

circuit which is used for most calibration work (see sec. 3.1.1).

Five (Vf-V-|o) comparisons were made using the ac calibrator voltage as the input signal. The mean values of these comparisons and the

standard deviation of a comparison for each frequency are shown in

table 4.

566
Table 4. Frequency response of the rms voltmeter with corrections

applied for variations in the ac calibrator voltage,

used for the input signal. The mean values and standard

deviation, Sf, were obtained from five ( V^-V

-j q) DVM

comparisons.

Frequency (f) (Vf-Vio)DVM Sf

5 Hz 0 ppm 6.1 ppm

2 -2 6.5

1 -13 4.5

0.5 -49 6.5

0.2 -109 5.5

0.1 -208 2.7

The voltmeter response as a function of input signal frequency can

be approximate d by sec ond degree equations of curves that pass through

the values of ( V^-V i o) D VM at frequencies of 2, 1, 0.5 Hz and at

0.5, 0.2, and 0.1 Hz. The equation that is applicable to the 0.5 to

2 Hz range is

(Vf -V 10 ) DVM = -105.33 + 133. OOf - 40.67f2

. (1)

The equation for the 0.5 to 0.1 Hz range is

( V

f -V 10)DVM

= - 346 - 50 + 1582. 50f - 1975. OOf 2

. (2)

These equations are used to plot the curves in figure 6.

As indicated previously, the lack of flatness in the DVM response

is attributed to the rms/dc converter response. Since applying

corrections from the curves in figure 6 or the preceding equations is

quite tedius, an effort was made to improve the DVM frequency response

sufficiently so that corrections usually are not necessary. This was accomplished by employing frequency-dependent negative feedback

circuits in a section of the input amplifier, as shown in figure 7.

With these circuits, the amplifier response approximately complements

the rms/dc converter response. Components Rl, CI, and R2, C2 provide

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high-frequency attenuations of approximately 250 and 30 ppm,

respectively. The corresponding corner frequencies are 0.14 and

0.71 Hz. The frequency response of the circuit which has the higher

attenuation is adjustable over a small range by the DVM FREQ RESPONSE

pot. Figure 8 is a plot of the amplifier transfer characteristic

with the DVM FREQ RESPONSE circuit switched in (switch S closed).

Resistances r-j and r? were arbitrarily assigned the value of 100 k^;

however, measurements show that the frequency response is essentially

independent of the values of r-| and selected by the V-RANGE-MV

switch (see fig. 2).

The frequency response of the circuit shown in figure 7 was combined with the responses of eqs. 1 and 2 and plotted in figure 9.

This response curve shows that the maximum values of ( Vf -V-j q )dvm occur at 0.5 Hz and approximately 0.14 Hz. A good estimate of the

outside limits of (Vf-V-jo)DVM can be obtained by measuring this

quantity at 0.1, 0.2, and 0.5 Hz, using the ac calibrator. Table 5 shows the DVM response at the lowest four calibrator frequencies,

obtained from 13 measurements. Corrections were made for the measured

frequency response of the calibrator (systematic error), leaving

only the imprecision of these measurements (±6.0 ppm) as the

uncertainty of the calibrator voltages. (The frequency response and

estimated standard deviation of the ac calibrator were based on 5 and 13 measurements, respectively - see section 3.1.1.) Using the

maximum estimated standard deviation of the measurement process

represented by table 5 (6.4 ppm) and a confidence level of 0.99, the

imprecision of these measurements is 5.4 ppm. Combining this

imprecision with the calibrator imprecision gives ±8.1 ppm as the maximum uncertainty of ( Vf-V-| q )DVM • Therefore, the outside

limits of (Vf-V-|o)DVM» measured at 0.1, 0.2, 0.5, and 1 Hz, are approximately +20 and -19 ppm.

Table 5. Measured DVM frequency response with input amplifier

compensated. The average values of ( Vf-V iq)dVM and

estimated standard deviations, Sf, are based on

13 measurements.

Frequency (f) ( vfv 10)DVM Sf

1 Hz +1 ppm 3.0 ppm

0.5 -11 4.2

0.2 +12 6.4

0.1 -7 5.0

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These limits result from using 5 and 13 measurements to establish

the frequency response of the calibrator and DVM, respectively. For

routine calibrations of the DVM frequency response, it is planned to

use three measurements. Assuming that the calibrator response is also

obtained from three measurements, the DVM and calibrator imprecisions

will be 11.3 and 7.8 ppm, respectively. Combining these quantities

gives 13.7 ppm, an increase of 5.6 ppm over the imprecision applicable

to the frequency response values in table 5. To obtain the maximum

expected values of (Vf-V"|o)DVM> using three DVM and three calibrator

measurements, the maximum frequency response values in figure 5 should be increased by 5.6 ppm, yielding +17.6 and -16.6 ppm. Adding

the corresponding imprecision to these quantities gives estimated

uncertainty limits of +31.3 and -30.3 ppm to ( Vf-V ]q)dvm* ^he

response curve in figure 9 indicates that (Vf-V-|o)ovM is approximately

4 ppm larger at 0.14 Hz than at 0.2 Hz. Also, any uncertainty in

the difference between ( V

^

-| 4-V ] q )qvm anc' ( ^ .2 0 )DVM 1n figure 6

carries over to figure 9. The computed value of this difference is

approximately 55 ppm. It is believed that the uncertainty in this

difference should not exceed 10 percent or 6 ppm. Adding this

uncertainty and the 4 ppm difference mentioned above to the uncertainty

limits for the measured frequency response yields +41.3 and -30.3 ppm

as the uncertainty limits of (Vf-Vio)DVM f° r an^ frequency. Since the

attenuation caused by the feedback circuits of the input amplifier

is small at 0.1 Hz but increases with frequency, the DVM response

curve (fig. 9) can be effectively rotated about the 0.1 Hz point of

the curve by adjusting the DVM FREQ RESPONSE pot. Therefore, this

control can be used to adjust the +41.3 and -30.3 limits to approximately

±36 ppm. Drift in ( V^-V -j q )dVM "that cannot be removed by adjusting

the DVM FREQ RESPONSE pot, estimated to be less than 2 ppm in five

years, can be compensated by trimming the values of R2 and C2 in

figure 7. However, to obviate the trimming, the frequency response

will be specified as 0 ±40 ppm to allow some margin for drift of

this type.

It is useful to estimate the uncertainty that the DVM response

causes if two voltages are compared which have frequencies differing

by no more than 5 percent. Figure 9 shows that the maximum rate of

change of (Vf-Vio)DVM w "" tln frequency occurs over the frequency range

of 0.1 Hz to 0.14 Hz, approximately. Assuming that (Vf-V]o)DVM changes

by no more than two-thirds of the specified limits of ±40 ppm over

this frequency range (a 40% change in frequency), a 5 percent uncertainty

in frequency corresponds to a maximum uncertainty in ( V-p-V iy )dvm °f

approximately ±7 ppm.

4. CALIBRATIONS OF VOLTMETERS

Calibrations are performed on rms-responding ac voltmeters using the

ac voltage calibrator of the NBS AC Voltmeter/ Cal i brator. AC voltmeters

can be calibrated at frequencies of 10, 5, 2, 1, 0.5, 0.2, and 0.1 Hz at

any voltage level between 0.5 mV and 7 V.

573
The total uncertainty of a calibration is ±( ec + er ), where

e c is the uncertainty of the ac voltage calibrator voltage and er

is the imprecision of the calibration process. The estimated uncertainty

of the ac voltage calibrator is ±0.020 percent, as described in section

3.5. The imprecision for a given calibration point is ts/M, where t

is Student's t for the selected confidence level, s is the standard

deviation for a measurement, and N is the number of measurements (test

runs) used to obtain the mean values of the measured quantities (voltmeter

readings ).

Unless otherwise requested by the customer, a single accuracy of

calibration will be given for the corrections to the calibration points,

based on a confidence level of 0.98 and the largest standard deviation

encountered. It is expected that the value of N, requested by the

customer, will generally be 3, 4, or 5. Assuming a selected confidence

level of 0.98, values of t corresponding to N equal 3, 4, and 5 are approximately 7.0, 4.5, and 3.7, respectively.

5. CALIBRATIONS OF VOLTAGE STANDARDS

Voltage calibrations are performed on precision ac sources using the

ac voltage calibrator and rms voltmeter of the NBS Voltmeter/ Cal i brator.

The rms voltmeter is used as a transfer voltmeter to compare the voltages

(of the same nominal level) from the test unit and the ac calibrator.

Voltage calibrations can be made at any frequency in the 10 Hz to 0.1 Hz

range and are usually made near the full-scale level of the DVM ranges

(2, 5, 10, 20, 50 mV, and 0.1, 0.2, 0.5, 1, 2, 5 V).

If the voltage standard is of a design which has inherently stable

voltage with respect to frequency, a frequency response (relative voltage)

calibration may be preferable to a conventional voltage calibration. The user then establishes the voltage levels by calibrating the voltage at

10 Hz. Since the rms voltmeter has a frequency response that is flat to

within ±40 ppm, it is used to determine the frequency response of

voltage sources. Frequency response calibrations can be made at any

frequency in the 10 Hz to 0.1 Hz range and are usually made near the

full-scale level of the DVM ranges (2, 5, 10, 20, 50 mV, and 0.1, 0.2,

0.5, 1, 2, 5, 10 V).

5.1 Voltage Calibrations

For each calibration point, the rms voltmeter (DVM) is used to

compare the voltage Vj from the standard under test with the voltage

V c of the same nominal level from the ac voltage calibrator, choosing

the calibrator frequency closest to that o f the vol tage being measured.

After the test runs are made, the average (T^ - Vj ) is formed for

each calibration point, yielding corrections to the nominal voltage

values of the test unit.

574
If the uncertainty of the calibrator voltage and the uncertainty

caused by the DVM frequency response are denoted by e c and em, respectively, the total uncertainty of a calibration point is given by

±(e c + e m

+ ts/M), where s is the standard deviation of a (V c

- Vj) comparison and N is the number of comparisons (test runs).

If the frequencies of the voltages being compared are within 5 percent

of each other, e m

is ±7 ppm. If the frequency difference is

larger, the worst case value of em, ±40 ppm, is used. The value

assigned to e c is ±200 ppm (±0.02%); however, if the highest

calibration accuracy is required by the customer, e c can be decreased

to ±125 ppm by a special calibration of the NBS instrument.

Unless otherwise requested by the customer, a single accuracy of

calibration will be given for the corrections to the calibration points,

based on a confidence level of 0.98 and the largest standard deviation

encountered. It is expected that the value of N, requested by the

customer, will be 3, 4, or 5.

5.2 Frequency Response Calibrations

These calibrations are made using the DVM to compare test unit

voltages of frequency f w ith the 10 Hz value. After N comparisons,

the quantities (Vf - V

-j q ) are computed for each test frequency, yielding

the frequency response. The uncertainty of these calibration points

is ±(e m

+ ts//N"), where e m

is ±40 ppm, the frequency response

of the DVM, and s is the standard deviation of a comparison.

Unless otherwise requested by the customer, a single accuracy of

calibration will be given for the frequency response, based on a confidence level of 0.98 and the largest standard deviation encountered.

It is expected that the value of N, requested by the customer,

will be 3, 4, or 5.

6. ACKNOWLEDGMENTS

The author is grateful for the valuable suggestions made by

Barry A. Bell, Group Leader for Electronic Instrumentation and

Metrology, and the CALC0M subcommittee members who reviewed this

document. The members of this subcommittee were: David W. Berning,

Ronald F. Dziuba, Dr. James A. Lechner, and John D. Ramboz. Special

thanks are due Mr. Ramboz, who served as chairman of the GAL COM

subcommittee, and Dr. Lechner, who provided assistance on the

statistical aspects of the measurements. The very helpful suggestions

by Jenny Pal 1 a, the typing of Bonnie L. Smith, and the editorial

assistance of Betty A. Oravec, in preparing this manuscript, are also

gratefully acknowledged.

575
7. REFERENCES

[1] AC Voltmeter/Calibrator Developed, NBS DIMENSIONS, Dec. 1977.

[2] H. K. Schoenwetter, "An RMS Digital Voltmeter/Calibrator for

Very Low Frequencies," IEEE Trans. Instrum. Meas., Vol. IM-27,

pp. 259-268, Sept. 1978.

[3] B. F. Field, "A Fast Response Low-Frequency Voltmeter," IEEE Trans.

Instrum. Meas., Vol. IM-27, pp 368-372, Dec. 1978.

[4] B. A. Bell, B. F. Field, and T. H. Kibalo, "A Fast Response Low

Frequency Voltmeter," Nat. Bur. Stand. (U.S.) Tech Note 1159

(Aug. 1982).

[5] H. K. Schoenwetter, "NBS Provides Voltage Calibration Service in

0.1 - 10 Hz Range Using AC Voltmeter/Calibrator," IEEE Trans.

Instrum. Meas.. Vol. IM-28, pp. 327-331, Dec. 1979.

[6] F. L. Hermach, "An Investigation of Multi junction Thermal Converters,"

IEEE Trans. Instrum. Meas., Vol. IM-25, pp. 524-528, Dec. 1976.

576
APPENDIX A

Calibration Forms

Several forms have been prepared to facilitate the calibrations of rms voltmeters and the voltage and frequency response calibrations of ac voltage standards. These are:

(1) LOG SHEET

(2) WORK SHEET

(3) TABLE OF CALIBRATIONS

(4) REPORT OF TEST

These forms, as well as a brief description of the calibration

procedure, are given in the following sections for each type of calibration.

Items 3 and 4 and the summarized calibration procedure are sent to the

customer, following a calibration.

577
Department of Commerce

VOLTMETER CALIBRATIONS USING NBS AC VOLTMETER/CALIBRATOR

National Bureau

of Standards Equipment Mfr.

LOG SHEET Submitted by:

Date Test No - Model No. Ser. No.

Time Frequency

Voltage

range Voltmeter readings (Vm ) Mean value

——

578
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579
Calibration of Voltmeters

Calibrations are performed on rms-respondi ng ac voltmeters using

the ac voltage calibrator of the NBS AC Voltmeter/ Cal i brator. AC

voltmeters can be calibrated at frequencies of 10, 5, 2, 1, 0.5, 0.2,

and 0.1 Hz at any voltage level between 0.5 mV and 7 V.

The total uncertainty of a calibration is ±(eC + er)» where

e c is the uncertainty of the ac calibrator voltage and ep is

the imprecision of the calibration process. The estimated maximum

uncertainty of the ac voltage calibrator is ±0.020 percent. The

imprecision for a given calibration point is ts//IT, where t is

Student's t for the selected confidence level, s is the standard

deviation for a measurement, and N is the number of measurements

(test runs) used to obtain the mean values of the measured quantities

(voltmeter readings).

Unless otherwise requested by the customer, a single accuracy of

calibration will be given for the corrections to the calibration

points, based on a confidence level of 0.98 and the largest standard

deviation encountered.

580
Department of Commerce

National Bureau

of Standards

TABLE OF CALIBRATIONS

Date Test No.

VOLTMETER CALIBRATIONS USING NBS AC VOLTMETER/CALIBRATOR

Equipment Mfr.

Submitted by:

Model No. Ser. No.

Frequency

Voltage

range

Average of voltmeter

readings

Vvm

Correction

to Vyn,

Number of readings

(N) Total uncertainty

Confidence

level used

for t

,

581
roRM NM-3U

(VN)

U.S. DEPARTMENT OF COMMERCE

NATIONAL BUREAU OF STANDARD*

WASHINGTON. D.C. 20294

REPORT OF TEST

LOW FREQUENCY VOLTMETER

Manufacturer

Model No.

Serial No.

Submitted by

This voltmeter was tested using the ac voltage calibrator of the NBS AC

Voltmeter/Calibrator at a room temperature of 23 ±1 °C.

The test points, average voltmeter readings, corrections, and total uncertainty

are shown in the attached TABLE OF CALIBRATIONS. Also shown is the confidence

level upon which the imprecision is based. The ac calibrator voltage

uncertainty is treated totally as a systematic error. Unless otherwise

indicated, the total uncertainty is based on the largest standard deviation

encountered for any of the test points. Measurements on the test unit were

made over a period of time which was too short to yield long-term stability

i nformati on.

For the Director

National Engineering Laboratory

Barry A. Bell, Group Leader

Electronic Instrumentation and Metrology

Electrosystems Division

Test No. 722/

Order No.

Date:

Attn:

582
CALIBRATIONS

OF

VOLTAGE

STANDARD

USING

NBS

AC

VOLTMETER/CALIBRATOR

Equipment

Mfr.

Model

No.

Ser.

No.

Submitted

by:

Mean

Calibrator

DVM

readings

Test

unit

Mean

DVM

readings

Department

of

Commerce

National

Bureau

of

Standards

LOG

SHEET

Date

Test

No.

Frequency

Calibrator

Test unit

Nominal voltage

Time

583
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584
Calibration of Voltage Standards

For each calibration point, the rms voltmeter (DVM) is used to

compare the voltage Vj from the standard under test with the voltage

V c of the same nominal level from the ac voltage calibrator, choosing

the calibrator frequency closest to that of the vol tage being measured.

After the test runs are made, the average (V

c

- Vy) is formed for

each calibration point, yielding corrections to the nominal voltage

values of the test unit.

If the uncertainty of the calibrator voltage and the uncertainty

caused by the DVM frequency response are denoted by e c and em, respectively, the total uncertainty of a calibration point is given

by ±(ec

+ e m+ ts//N), where s is the standard deviation of a (V c

- Vj) comparison and N is the number of comparisons (test runs).

If the frequencies of the voltages being compared are within

5 percent of each other, e m

is ±7 ppm. If the frequency difference

is larger, the worst case value of em> ±40 ppm, is used. The

value assigned to e m is ±200 ppm (±0.02%); however, if the highest

calibration accuracy is required by the customer, e c can be decreased

to ±125 ppm by a special calibration of the NBS instrument.

Unless otherwise requested by the customer, a single accuracy of

calibration will be given for the corrections to the calibration

points, based on a confidence level of 0.98 and the largest standard

deviation encountered.

585
Department of Commerce CALIBRATIONS OF VOLTAGE STANDARD USING NBS

National Bureau AC VOLTMETER/CALiBRATOR

of Standards

Equipment Mfr.

TABLE OF CALIBRATIONS

Submitted by:

Date Test No.

Model No. Ser. No.

Nominal voltage of standard

Frequency of standard

Correction to

nominal voltage

Number of comparisons

(N) Total uncertainty

Confidence

level u^pri for t

—. ,

586
fonmMM^U

U.S. DEPARTMENT OF COMMERCE

NATIONAL BUREAU OF STANDARDS

WASHINGTON. O.C. 20X94

REPORT OF TEST

LOW FREQUENCY VOLTAGE STANDARD

Manufacturer

Model No.

~

Serial No.

Submitted by

This voltage standard was tested using the ac voltage calibrator and rms voltmeter (DVM) of the NBS AC Voltmeter/Calibrator at a room temperature of

23 ±1°C.

The test points (nominal voltage and frequency), corrections, and total

uncertainty are shown in the attached TABLE OF CALIBRATIONS. Also shown

is the confidence level upon which the imprecision is based. The uncertainty

of the ac calibrator voltage and the uncertainty caused by the DVM frequency

response are treated as systematic errors. Unless otherwise indicated, the

total uncertainty is based on the largest standard deviation encountered for

any of the test points. Measurements on the test unit were made over a period

of time which was too short to yield long-term stability information.

For the Di rector

National Engineering Laboratory

Barry A. Bell, Group Leader

Electronic Instrumentation and Metrology

Elect rosy stems Division

Test No. 722/

Order No.

Date

Attn:

587
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Frequency Response Calibrations

These calibrations are made using the DVM to compare test unit

voltages of freq uency f w ith the 10 Hz value. After N comparisons,

the quantities fVf - V-j q ) are computed for each test frequency,

yielding the frequency response. The uncertainty of these calibration

points is ±(e m+ ts//N), where e m

is ±40 ppm, the frequency response

of the DVM, and s is the standard deviation of a comparison.

Unless otherwise requested by the customer, a single accuracy of

calibration will be given for the frequency response, based on a confidence level of 0.98 and the largest standard deviation encountered.

590
Department of Commerce FREQUENCY RESPONSE CALIBRATIONS OF VOLTAGE STANDARD

National Bureau USING NBS AC VOLTMETER/CALIBRATOR

of Standards

Equipment Mfr.

TABLE OF CALIBRATIONS

Submitted by:

Date Test No.

Model No. Ser. No.

Nominal

voltage Frequency

Frequency

response

( vf- v io)

Number of

comparisons

(N)

Total

uncertai nty

Conf 1 dence

level used

far t
FMUNU4U

U.S. DEPARTMENT OF COMMERCE

NATIONAL BUREAU OF STANDARDS

WASHINGTON, D.C XOtM

REPORT OF TEST

LOW FREQUENCY VOLTAGE STANDARD

Manufacturer

Model No.

Serial No.

Submitted by

This voltage standard was tested for frequency response using the rms voltmeter (DVM) of the NBS AC Voltmeter/Calibrator at a room temperature

of 23 ±1°C.

The test points (nominal voltage and frequency), corrections, and total

uncertainty are shown in the attached TABLE OF CALIBRATIONS. Also shown

is the confidence level upon which the imprecision is based. The uncertainty

caused by the DVM frequency response is treated as a systematic error. Unless otherwise indicated, the total uncertainty is based on the largest

standard deviation encountered for any of the test points. Measurements

on the test unit were made over a period of time which was too short to

yield long-term stability information.

For the Director

National Engineering Laboratory

Barry A. Bell, Group Leader

Electronic Instrumentation and Metrology

Elect rosy stems Division

Test No. 722/

Order No.

Date

Attn:

592
APPENDIX B

Detailed Design of Voltage Calibrator

593
594
595
Description of Components Used in Voltage Calibrator

Amplifiers AR1 and AR2 are Analog Devices Type 52K.1

Capacitors CI and C2 are 5 percent silver mica types. Capacitor

C2, or associated timing components, may need trimming to yield the

desired clock frequency. The capacitors associated with AR2 should

be trimmed so that (capacitance) (cal ibrator frequency) = constant,

to within ±0.2 percent. Nominal values (pF) are: C

a

= 0.025,

C b

= 0.05, C c= 0.125, C d= 0.25, Ce

= 0.5, Cf = 1.25, and C

g

= 2.5.

These polycarbonate capacitors have a rating of 100 V, 5% tolerance

and ±0.1%/°C temperature coefficient. They were shunted with

smaller capacitors (mica or polycarbonate) to obtain the desired

capacitance ratios.

The function generator (U8) is Micro Networks Corporation Type

MN350.

The dc reference ( PS 1 ) should have a dc stability of better than

±5 ppm/year.

Resistors R12 and R13 are 1W precision wirewound resistors with

±0.01% tolerance and temperature coefficients matched to within 2 ppm/°C.

Two parallel 100 q resistors may be used for R13. All trimpots are

1W, 25 turn, panel mount type with ±150 ppm/°C temperature coefficient.

Resistors Rl - R4, R8, R14 - R16, R23 - R25 and R28 - R30 are Vishay

Resistive Systems Group Type 51 02 C. A non-inductive wirewound resistor

was used for R17. R27 is a sensistor from Texas Instruments, whose

value is determined from temperature tests. R7 is a general resistance

Type DV-4007A K-V divider.

Switches SI - S4 should have plastic sleeves on their handles to

minimize thermal voltage generation at the switch contacts.

1 Certain commercial equipment, instruments or materials are identified

in this report in order to adequately specify procedures or special

circuits. In no case does such identification imply recommendation or endorsement by the National Bureau of Standards, nor does it imply

that the material or equipmment identified is necessarily the most

suitable for the purpose.

596
APPENDIX C

Shipping and Fee Information

General shipping instructions are given in the latest issue of

NBS Special Publication 250, "Calibration and Related Measurement

Services of the National Bureau of Standards." Shipments of voltmeters

and voltage sources 'for calibrations should be directed to

National Bureau of Standards

Electrosystems Division, MET B162

Rt. 270 and Quince Orchard Road

Gaithersburg, MD 20878 .

Current information on calibration fees is given in the NBS Special

Publication 250 Appendix, which is updated every six months.

597
THE PRACTICAL USES OF AC-DC TRANSFER INSTRUMENTS

Earl S. Williams

ABSTRACT

Alternating currents and voltages are measured most accurately at this time

when they are compared with nominally equal and known dc currents and voltages.

The comparisons are usually made with thermal transfer instruments which

respond nearly equally to ac and dc quantities. Practical information and

recommended procedures are given for using these instruments along with

diagrams of apparatus and examples of typical data and calculations. Methods

for minimizing difficulties caused by dc reversal differences, thermal drift,

energy picked up from local electromagnetic fields, and the deviation from

square-law response of these instruments are considered. Causes of ac-dc

differences are discussed, and methods for measuring them and applying corrections are also described.

Key words: ac current measurements; ac voltage measurements; ac-dc comparator; ac-dc difference; thermoelement.

1.0 Introduction

The basic units for electrical measurements are derived at the National Bureau

of Standards in absolute terms from the units of mass, length, and time. The

standards primarily used to transfer these units for direct current measurements from NBS to other users, and to preserve them in the standards laboratories, are the saturated standard cell (the volt) and the standard

resistor. With these and a variety of shunts and resistance ratio instruments

for range extension, direct current and voltage can be measured with high

accuracy.

Since there is no ac counterpart to the standard cell, measurements of

alternating current, voltage, and power are made relative to these same standards. The chain of measurements is extended to these alternating quantities by ac-dc transfer instruments, which have a flat and known frequency response and hence may be calibrated on direct current and then used

for alternating current measurements. They provide an accurate transfer from

direct voltage and current standards to alternating current and voltage

measurements, hence the name.

Electrodynamic instruments have been used to transfer voltage, current, and

power; however, they are limited in frequency to a few thousand hertz due to

their relatively high inductance. Electrostatic moving systems are usable to

higher frequencies, but they have severe low voltage limitations. Instruments

which respond to peak and average values of ac voltage have been employed

successfully over certain voltage and frequency ranges (see section 3.8). The

instruments most widely used for voltage and current transfer measurements,

and the ones to be discussed here, are electrothermal. They are usually

called thermal voltage converters (TVCs) and thermal current converters (TCCs)

to distinguish them from other types of transfer instruments [1].

598
2.0 Thermoelements

The key component in these transfer instruments is a thermoelement (TE),

consisting of a heater which carries the current to be measured and a thermocouple attached to its midpoint. The output from the thermocouple is relatively low (7 to 12 mV); however, heater current differences, and differences

between ac and dc currents, as small as one part-per-mi 11 ion (ppm) can be

detected, if changes in the output are monitored with a sensitive detector--a

galvanometer or microvoltmeter. Thermoelements for currents from 1 to 1000 mA

are available with evacuated glass-bulb enclosures and with insulation (a

small ceramic bead) between the thermocouple and heater (fig. 1). TEs are available for currents up to at least 20 amperes, but vacuum enclosures are

not feasible at these currents because the elements require heavy copper

conductors which are very difficult to pass through a vacuum seal. Insulation

is not generally available in TEs for currents greater than one ampere. They

are difficult to construct, and the demand for them is not great. Most

current comparisons are made with special ac shunts in parallel with lowrange TEs (see section 8).

Figure 1. Vacuum thermoelement. (Heater and thermocouple wires retouched.)

The TE heater is short and straight thus having minimal reactance, and the

insulation of the thermoelement prevents any appreciable interaction between

the ac current and the read-out instrument. TEs are therefore well suited

to measurements at audio and higher frequencies. They are used with series

resistors in multi range TVCs up to 100 kHz on most ranges and to 1 or 2 MHz at

low voltages. Single-range units with carbon or metal film resistors coaxial ly

mounted in tubular casings are used up to 100 MHz.

Multi junction thermoelements [2,3,4,5], with a relatively low heater

temperature and 50 to 200 thermocouples in series distributed evenly along the

heater, are capable of higher accuracy in transfer measurements, and their

structure minimizes most of the thermoelectric effects that cause errors in

599
other types of TEs. However, they are difficult to construct, and therefore

expensive and not readily available.

2.1 Output Characteristics

The thermocouple output of the TE is a function of the power dissipated

in the heater, and therefore varies approximately as the square of the heater

current. However, the device does deviate significantly from a square-law

response as the heater current approaches the rated value. The relationship

of the output, E, to the heater current, I, may be expressed as

E = kl

n

. (1)

The response appears to be nearly square-law (i.e., n=2) at very low currents,

but n is usually 1.6 to 1.9 at rated heater current. The factor k varies

somewhat with large changes in heater current, but this is not a constraint.

The TE is a substitution device, and not normally used to measure ac current

as a function of thermocouple output. The factor k is constant over a narrow

range where nearly equal ac and dc currents are compared.

The relationship between a small change in TE heater current (AI) and the

corresponding change in output (AE) is expressed as

AI/I = AE/nE. (2)

If the thermoelement is in a TVC,

AV/V = AE/nE. (3)

It is often advantageous to make measurements of n and use the data in

calculations as suggested in sections 3.2 and 6.2. The measurements are not

difficult, and the values are quite permanent. Methods for measuring n are described in section 7.

2.2 DC Reversal Difference

DC reversal difference is generally (and in this document) defined as the

percentage difference between the dc current and its reversed counterpart,

when they both produce the same output from a TE. This difference is not

necessarily constant and may increase in some TEs as the heater current is

lowered. A low dc reversal difference is, of course, advantageous. The

difference between ac and dc voltage and current is often measured by

observing the change in indication of an instrument which responds to changes

in TE output as ac and dc are applied. If the dc reversal difference is

large, it may be necessary to reduce the sensitivity of the read-out instrument to keep all readings on scale.

It is recommended that both directions of dc be used in the procedures

described here. However, it is not often necessary to have an explicit

measurement of dc reversal difference. A measurement may be desirable

sometimes, as for instance when making sure that new TEs conform to specifications. A low- range TE can be connected in series with a resistance of 50 to

200 kfi, and the appropriate voltage applied for rated heater current. The

value of the two direct voltages which give equal output can be read from a

600
calibrated source or measured with a digital voltmeter. Common currentmeasuring equipment, usually a potentiometer or a digital voltmeter and a shunt, is used to measure currents in higher range TEs. In any method, it is

advisable to use a test sequence which will minimize the effect of drift in

the TE output. Readings should be taken with DC+, DC-, DC+ after time

intervals which are about equal. The average of values obtained with the two

positive dc voltages should be used in the calculations.

DC reversal difference can also be measured conveniently with a second or reference TE of about the same range and with known n values. The two heaters

are connected in series through a switch which reverses the test TE input but

not that of the reference one (fig. 2). Direct and reversed currents are adjusted for equal output from the test TE (null on detector Dl). The emf of

the reference TE (E) is read from its potentiometer (Pot'r) and changes in emf

(AE) are computed from readings taken from the detector (D2), which must be a microvol tmeter or a calibrated galvanometer. DC reversal difference, AI/I, is

then computed from eq. (2).

Reference TE

DC+ DC

DC Current

Supply Q$Z Pot' D1

Test

TE

Figure 2. Circuits for dc reversal difference measurements.

As pointed out in the previous section, a TE output varies approximately as the square of the heater current. If dc reversal difference were to be

defined as the percentage difference between emfs obtained with equal currents

in opposite directions, the figure would be about twice as large.

DC reversal differences range from near zero to several hundredths of a percent. They are largely due to Peltier heating and cooling at the ends of

the heater, and to Thomson heating in the two halves of the heater [6,7].

Both of these effects cause asymmetry in the temperature rise of the heater.

The asymmetry is affected when the current is reversed, and therefore the

temperature at the point where the thermocouple is attached is changed. DC

reversal difference is minimized by placing the thermocouple at the thermal

center of the heater. If the heater wire is not quite uniform, this may not

be the exact mechanical center. DC reversal difference is therefore not

easily controlled. Some manufacturers sort TEs according to dc reversal

difference and market those with low dc reversal difference as a premium

product.

601
DC reversal difference rarely changes with use, and the few changes that

have been observed are believed to be due to the heater having been overloaded. One possible explanation for this is that one-half of the heater was stressed more than the other, and its thermal center was shifted relative to

the thermocouple's position. If this model is correct, the dc reversal

difference could be either increased or decreased by an overload.

i 2.3 Temperature Coefficient

The temperature coefficient of most TEs (percentage change in output for a temperature change of 1 °C with constant heater current) is relatively

high—often 0.1%/°C. Elements should therefore be mounted in a thermally

lagged enclosure to minimize the effect of ambient temperature changes. The

output of a TE can be very stable at a fixed temperature if the heater is

carrying a constant current for several hours. However, if the instrument is

switched back and forth between ac and dc after a brief warm up, as it is in

most measurements, the output will usually drift. The drift may be negligible

during the short time required for a measurement, especially at low voltages.

However, on many ranges where there is some self- heating effect, the drift

rate is apt to be quite constant but significant. Accurate and consistent

measurements can be made under these circumstances, if readings are taken at

time intervals that are about equal and in a sequence which tends to average

out the effect of drift (e.g., [a] +DC, AC, -DC or [b] AC, +DC, -DC, AC).

2.4 AC-DC Difference

Thermoelements, as well as TVCs and TCCs, sometimes have a significant ac-dc

difference, 6, particularly at higher frequencies,

« (v, - y/vd

,

where V and V . are the ac voltage and the average of the two directions of dc

voltage required for equal response, or output emf. For voltage measurements,

V

a = Vd (1 6). (4)

Similarily, for current converters, I g = 1^ (1+6).

AC-dc differences, methods for measuring them, and the application of

corrections are discussed in sections 5.0 to 6.5.

2.5 Radio Frequency Interference

Thermoelements have a frequency response which extends to 100 MHz or more, and

measurements made with them can therefore be affected by local electromagnetic

fields. Interference from television stations can be particularly troublesome

with low-current TEs, because the length of leads used in calibration laboratories often makes them an effective antenna. Energy picked up by a measure- ment circuit can usually be detected by shorting the TVC input terminals with

all connections in place as in normal use, but with the power supplies off.

Any change in the indication of a detector in the TVC output circuit as the

short is opened and closed will indicate a pick up problem. Interference can usually be avoided by using coaxial leads and shielded circuit components. In

extreme cases it may be necessary to work in a shielded room.

602
3.0 Voltage Measurements

AC voltage is measured most accurately at this time by comparison with a nominally equal dc reference voltage using a thermal voltage converter. The

TVC may consist simply of a series resistor and a thermoelement mounted

coaxial ly in a tubular casing, usually a plated brass tube, with input and

output connectors [8]. As noted earlier, this configuration is well suited to

rf measurements. In another arrangement, the resistor is mounted in one tube

and the TE in another. The TE can then be attached to any one of a set of

resistors to make a TVC with the desired voltage range [9]. The number of

resistors in a set can be reduced to about half, by using two TEs with different current ratings (e.g., 2.5 and 5 mA). Each resistor is then used for

two ranges [10]. These sets, which are commercially available, have two

distinct advantages. The ac-dc difference corrections are very small, and

they can be determined relative to any one range, which has known corrections,

by a step-up or step-down intercomparison of adjacent ranges.

Multi range TVCs usually contain 10 to 14 ranges between 0.5 and 1000 volts,

and ranges are selected with a rotary switch which connects the TE in series

with one of the ranging resistors. Some commercial models also incorporate an ac-dc switch, as well a null detector, and a balancing emf source for

monitoring the TE output. (The "balancing emf" referred to here and elsewhere

usually consists of the voltage across a fixed resistor with current supplied

by a battery and adjusted with a resistive divider or series resistors. The

circuit may be quite similar to the Lindeck potentiometer shown in fig. 3, but

without the mi 1 1 iammeter. It balances, but does not measure, the emf.) If

these components are not included in the TVC, the relatively simple Lindeck

potentiometer (fig. 3) can be used with a detector, which may be either a microvol tmeter or a galvanometer, to monitor the TE output.

The detector can be simply used as a null detector when an ac voltage is

adjusted to be equal to a reference dc voltage (3.1), or a calibrated detector

can be used to measure the changes in TVC output emf as ac and dc voltages are applied alternately. The difference between the ac voltage and the dc

reference is then computed from the emf differences (3.2). Provisions are

made, and instructions furnished, by some manufacturers for using either

method.

3.1 A Null Method

In the "null" procedure, the ac voltage to be measured is applied to the TVC

and the balancing emf is adjusted for a detector null. The two polarities of

dc voltage are then applied in turn, and each one is adjusted to produce a null indication on the detector. The two voltages are measured (or read from

a calibrated source), corrected, and averaged. The average is corrected for

the ac-dc difference, 6, of the TVC to give the RMS amplitude of the ac voltage. For example, from eq. (4), with 6 = -0.012%, V.+ = 200.012 V, and

V

d

- = 199.960 V,

V = [(200.012 + 199.960)/2] + (V. x -0.012/100)

a = 199.986 - 0.024

= 199.962 V.

603
AC voltage should be reapplied after the two dc voltage measurements, to

evaluate any significant drift from the original null position. To find the

magnitude of the drift, the detector can be calibrated by making a small change

in the dc voltage, observing the change in detector indication, and computing

the scale sensitivity in percent per division. If the drift is large, a longer

warm up period should be allowed, or the "deflection method" (3.2) may be used.

In the example above, a small part of the 0.026 percent difference between the

two dc voltages may be due to drift in the TVC output, but most of it is due

to the dc reversal difference. These measurements are sometimes carried out

with only one direction of dc voltage, and a correction equal to half the dc

reversal difference is then applied. This must be done very cautiously. If

the correction is applied with the wrong sign, the error is twice as large as with no correction. Of course, if the dc reversal difference is small relative

to the desired accuracy, satisfactory measurements may be made using only one direction of dc current flow.

Figure 3. AC-dc switching, TVC and potentiometer for ac-dc comparisons.

NOTES:

The "DC Supply" may be a calibrator and serve as both source and standard,

or the voltage may be measured with a "DC Std. "--a digital voltmeter or a volt

box and a potentiometer.

R4 and R5 (see section 3.3) and R6 and R7 (section 3.6) are optional. They

are used to adjust and measure the detector sensitivity, respectively.

604
The null procedure has certain disadvantages which should be kept in mind:

(A) If the dc standard is one of the widely used five or six dial calibrators,

adjusting it for a detector null may be tedious and time consuming; (B) the

wear on the lower two or three control switches will be accelerated; (C) the

calibration of the dc standard may be in some question if a number of readings

are obtained above and below the round-number value where a calibration test

was made; (D) if the TVC output drift is significant during the time required

for the null adjustment, it may be difficult to obtain consistent results.

Where the drift rate is constant, and the measurements are made in equal time

spans, the results may be consistent but in error; and (E) if the dc voltage

is supplied by a power supply and measured with a volt box and potentiometer,

the time required for the two adjustments (the voltage adjustment for a TVC

null and then the potentiometer balance) may be even longer. The two adjustments can be made in less time if two operators work together.

3.2 A Deflection Method

A calibrated detector may be used with a TVC to measure the difference between

a dc reference voltage and an ac voltage. The difference can be applied as a correction, C, to the dc voltage to obtain the value of the ac voltage.

V

a = Vd (1 + C) (4A)

(see section 3.5, Calculations)

This procedure will largely avoid the problems listed above. It should also

save time, as the adjustments are fewer and less tedious. The data and

calculations may be a bit more complicated; however, once routine procedures

are established, this method can be used with confidence. If the correctness

of the results is in question, they can be verified by relatively simple extra

tests, as described in section 3.5.

In the deflection method, the dc voltage is adjusted to a round-number value

which is nominally equal to the ac voltage to be measured. The TVC is

connected to the dc voltage (see fig. 3), and the balancing emf (or

potentiometer) is adjusted for a near null on the detector. The TVC is then

switched to the ac and dc voltages in the sequence indicated in fig. 3 (DC+,

AC, DC-), and the small emf variations are read from the detector directly or computed from the detector readings,* as explained below. The difference

between the ac and dc voltages is computed from these readings.

An equation very much like (3) in section 2.1 is used. Instead of a simple

change in TVC input voltage (AV/V), the difference between the ac voltage (V

g

) and the corrected dc voltage (V

d

) is measured by observing the change in TVC

output. Therefore, AE in eqs. (2) and (3) becomes E &

- E d>

the emfs

corresponding to the voltages V and V

d

- The eq. (4A) above may therefore be

written

* The detector should be connected so the indication increases, or deflects up

scale, when the input voltage to the TVC is increased.

605
v

a = vd

(i + c>-vd

(i + +

where, as before, n is the exponent in eq. 1, E is the output of the thermoelement, and 6 is the ac-dc difference correction to the TVC.

The emfs E

fl and E d are the TVC outputs (EQ

, fig. 3) with ac voltage and the

two directions of dc voltage applied to the TVC. However, all but a small

fraction of these emfs are balanced by the emf source or the potentiometer.

The difference between the two emfs can be computed from imbalance emfs, e, read from a microvol tmeter.

The procedure is probably most straightforward when a digital microvol tmeter

is used. Emfs are indicated directly in volts (pV or nV). AC and dc

voltages should be applied, and emf readings taken, in time intervals which

are about equal. Apply DC+, AC, and DC-, read e lf e 2 , and e3 , and compute

E a

• Ed = e

2

" < e

l + e

3 )/2 '

The complete calculation is discussed in section 3.5. The digital microvoltmeter indicates both positive and negative readings with the sign displayed, but mistakes are less likely to occur if the balancing emf is adjusted

so that all readings have a positive sign.

Emfs may also be read directly from a deflecting microvol tmeter. However, it

is recommended that readings be made in divisions, a deflection change, AD,

computed, and the emf difference E - E . be computed from the deflection

a q

change. Techniques for reading detector scales are described in section 3.7,

and multiplying factors, K, for converting AD to emf differences are also

discussed. Calculations from data taken by this procedure are shown in table

1, column 5, where

E

a

" E d = (D

2

" (D

1

+ °3 )/2) x K '

Emf differences may also be measured by a calibrated galvanometer.

3.3 Electronic Galvanometers

An electronic galvanometer with appropriate sensitivity may be used to measure

an emf difference, if a method is provided to determine its voltage sensitivity, K, as described in section 3.6. Electronic galvanometers are available

with continuously adjustable gain control over a wide range, and this feature

provides a significant advantage. The sensitivity can be adjusted for maximum

resolution for each measurement. On the other hand, a deflecting microvoltmeter's sensitivity drops typically by about 60 percent if the range is

increased (e.g., from 10 to 30) to keep all readings on scale.

606
Table 1. Examples of Calculations

In these examples, data are listed in columns 3 and 5 and identified in

columns 1, 2, and 4. Test parameters are listed in the first eight lines. An

emf difference, ave AE, is computed from emf readings e^ to e^ in column 3 and

from deflection readings to Dp. in column 5. See text for further explanation.

1 2 3 4 5

Freq kHz 50 Freq 50

Range test inst. 200 Range 200

v

d

DC voltage(V) 200.006 v

d

200. 006

TVC i cs 1 1 y c TVC 200

E TVC emf (mV) 8. 8 E 8.8

n 1. 83 n 1.83

nt 16. 1 nt 16. 1

pvm range 10 uVm 10

K 2 x 10" 4

e l

dc 6. 5 D

i

23

e

2 ac 4. 4 D

2

36

e

3

dc (rev) 7. 1 D

3

73

e4 ac 4. 1 D

4

35

e

5

dc 5. 8 D

5

20

AE

1

* uV -2. 4 AD

1

-12

AE

2

* uV -2. 3 AD

2

-11

ave AE mV -0. 0024 ave AE -0.0024

100 AE

percent -0. 015

100 AE -0.015

nE nE

6 TVC corr % +0. 003 6 +0.003

C Final corr % -0. 012 C -0.012

Notes: *AE

X = e2

- (e

1 + e

3 )/2 and AE

£ = e4

- (e

3 + e 5 )/2

e 1

and e

& differ because of drift in the TVC output. e

1

and e

3

differ partly

because of drift but mainly due to dc reversal difference in the TVC. In the

statements above, substitute D for e where the readings are in divisions.

607
If a "one-range" instrument intended mainly as a null detector is used, an arrangement to adjust its sensitivity is often desirable. Such detectors can

be shunted by resistors, such as R 4 and R 5 in fig. 3. When the potentiometer

is balanced, the detector responds to currents through these resistors. The

switch, which must have contacts with very low noise and thermal emfs, may

select one or several shunting resistors. The resistor values should be

chosen experimentally, depending on the basic sensitivity of the instrument

and the sensitivity desired. Difference ratios of 2/1 or 3/1 between

sensitivities are recommended. These instruments may be calibrated as suggested in section 3.6, and the scale reading techniques of section 3.7 are

useful also.

3.4 D'Arsonval Galvanometers

D'Arsonval (permanent magnet, moving coil) galvanometers are often considered

to be quaint, if not obsolete, by electronically oriented metrologists , and

buying one for these measurements is not recommended. However, if an appropriate galvanometer is readily available, as is often the case, it should not

be rejected out of hand. It is probably superior to most electronic instruments in two important respects—noise rejection and zero stability. Its

sensitivity will probably be quite low unless a photoelectric galvanometer

amplifier is used to increase its input voltage. However, some galvanometers

may be sufficient for certain measurements where the voltage differences are

large, or where there is less need for accuracy.

It is advisable to test a galvanometer for linearity if a deflection method is

to be used by applying small equal changes to the potentiometer input and

seeing that the resulting changes in indication are about equal on all

portions of the scale. Some instruments may be nonlinear at the extreme ends,

but satisfactory over most of the middle portion.

3.5 Calculations

Table 1 is intended to illustrate data logging and calculations and to suggest

a data sheet format. Different data arrangements and line labels may be

preferred by some users, but it is advisable to develop a more or less standardized data sheet for each type of measurement. The line labels in column 1 are brief to conserve space (E - E . is abbreviated to AE as in eqs. (2) and

(3)), but they should be sufficient for a practical data sheet. Labels are further explained or defined in column 2.

In column 3, a correction, C, is determined for a test instrument indicating

200.000 V at 50 kHz. The measurements are made relative to a corrected dc

reference voltage of 200.006 V. Two emf differences, AEj and AE2 , are computed from five readings of a digital microvol tmeter. The average AE is

changed to millivolts to be in the same units as the TVC output, E, before

being divided by nE. (Measurements and plots of n are discussed in section

7.0.) AE/nE is multiplied by 100 to obtain a percentage value, and the ac-dc

difference correction to the TVC (6) is added to determine the correction, C,

to be applied to the dc voltage to obtain the ac value.

608
The same test is repeated in columns 4 and 5, but the data are read in

divisions from a deflecting microvol tmeter. The line labels in columns 1 and

4 are very similar. Deflections D x to D 5 are read rather than emfs, ADj and

AD 2 are computed, and the average is multiplied by conversion factor K = 2 x

10 -4 (from table 2) to find the average AE in millivolts. The rest of the

calculation is the same as in column 3. If data in ppm are preferred, the

average AD can be multiplied by K=200 from table 2 to obtain -2400 nV. Then

AE/nE + 6 = -2400/16.1 + 30 = -120 ppm .

The correction, C, expressed in percent, can be applied to the reference

voltage, V^, in two ways, as illustrated below, to obtain the measured value

of ac voltage, V . The calculation on the right may be preferable where a

Q

machine calculator is not used.

V

a = Vd (1 + c/10°) Va = V

d + ( V100 x C) = 200.006 (1 + (0.01 x -0.012)) = 200.006 + (2 x -0.012)

= 200.006 (.99988) = 200.006 - 0.024

= 199.982 V = 199.982 V

where C is expressed in percent

*** If the accuracy of the procedure or the correctness

of the sign is in question, it can be verified as follows: Increase the dc voltage by 0.020 percent after

a measurement, and repeat the procedure and calculation.

The correction, C, should be 0.020 percent more NEGATIVE

than in the first measurement.

3.6 Galvanometer Calibration

An obvious method for calibrating a galvanometer scale in these test circuits

is to introduce a small measured change in the TVC input, observe the resulting change in indication, and compute a multiplying factor, K, for converting

changes in indication to voltage differences (e.g., percent/div or ppm/mm).

This method has certain advantages. No n data are required, and the TVC

output, E, need not be measured. It is also quite feasible in voltage measure

ments if the small voltage change can be measured with the dc standards being

used. The change should be large enough for good resolution, but of course,

the galvanometer must not deflect off scale.

This calibration method is, however, more tedious and time consuming than some alternate methods. It may also require additional equipment in ac-dc difference measurements (see 6.2) where no dc standards are ordinarily used.

Several calibration methods have been developed to introduce a known change in

emf directly in the detector circuit. In one of these, a change in resistance

(AR) in the balancing emf circuit produces a change in current and hence a change in emf. However, this requires a rather sophisticated circuit with

practically constant resistance (except for the AR) at all emf levels.

609
A relatively uncomplicated calibration circuit, which requires its own battery,

is shown in fig. 3. It consists of R6 and R 7 , an on-off switch and a 2400-

mAh mercury battery. A computable emf, e, is inserted across R 7 (a one percent, 1-ohm resistor is recommended) by closing the switch. A round-number

emf of 10 (jV (R6=135 kfi) will be satisfactory in many instances; however, the

calibrating emf should be commensurate with the detector sensitivity. A

rotary switch with several contacts can be used to connect any one of several

resistors in the R6 position. It may be desirable to have more than one calibrating emf if the sensitivity of the detector is adjustable over a wide

range.

A multiplying factor, K, (uV/div) is computed from the deflection observed

when the voltage is inserted. The deflection change, AD, (sections 3.2, 6.2

and 8.0) resulting from differences between ac and dc voltages or currents, is

multiplied by K to obtain the emf difference, E - E.. This factor K is

a u equivalent to K in table 1 (column 5). But in that example the detector is a microvol tmeter, and K is obtained from table 2.

Table 2. Conversion Factors K

Microvolts Divisions u_V K (x by) for 100-div scale

uV (full (full div to div to div to

range scale) scale) millivolts microvolts nanovolts

30 60 60 6 x 10~l 0.6 600

10 100 20 2 x 10~Z 0.2 200

3 60 6 6 x 10 I 0.06 60

1 100 2 2 x 10

3 0.02 20

The current drawn by the battery is, of course, minimal, and the mercury cell

maintains a nearly constant voltage for most of its useful life. A new battery should last a year unless it is inadvertently left on for an extended

period. It is advisable to check its voltage occasionally by connecting a voltmeter between the positive input connection and the test point "P" (fig.

3). The potentiometer switch may be either on or off.

3.7 Reading Deflecting Detectors

The scales on most suitable microvol tmeters and galvanometers have the zero indication at the center, so that positive and negative readings can be made

to the right and left. Microvol tmeters with intermediate ranges —3, 30, 300,

etc. —usually have scales with 30 divisions to either side of zero as well as 50-division scales for the decade ranges. This arrangement is convenient for

most measurements and for null detectors. However, it is very desirable to

have the zero at the far left for the procedures described here. Typical

microvol tmeter scales are shown in fig. 4, but with a number plate in front of

the 50-division scale. The substitute numbers are engraved on a plastic plate

and filled with black wax. A similar plate can be made by cementing cut-out

numbers on a transparent plate. The instrument is read with zero at the left

and a full scale range of 100 divisions. It can be read in microvolts;

however, some conversion to other units will probably be advantageous in the

610
calculations. It is probably more straightforward, and mistakes are less

likely, if readings are recorded in divisions. A deflection difference, AD, can be computed in divisions and then converted to an emf difference, AE,

using the appropriate factor, K, from table 2 below.

Figure 4. Typical microvol tmeter scales with three deflections marked.

Three readings are marked on the scale in fig. 4 at 23, 36, and 73 divisions

as an illustration. If the reading at 36 divisions corresponds to an ac voltage or current and the others to direct and reversed-dc, then

AD = D

2

- (D

1 + D3 )/2 = 36 - (23 + 73)/2 = -12 div.

Of course, the same result is obtained when the regular scale is read, but the

sign must then be observed very carefully. The readings are -27, -14, and

+23, and

AD = -14 - (-27 + 23)/2 = -12 div.

Obviously, mistakes are less likely with the substitute numbers, where all

readings are positive.

The scales shown in fig. 4 are typical of several instruments in which the zero and full scale indications on either scale coincide. It is feasible- -and

advantageous—to read the 100-division scale with any range on most of these

instruments. The same substitute numbers can be used with any range, and the

reading precision is better with 100 divisions. The feasibility of this

procedure must, of course, be verified for any particular instrument. In the

example in fig. 4, the indications marked on the 100-division scale are extended to the 60-division scale and read as 14.0, 21.5, and 43.5. Then AD =

21.5 - (14 + 43.5)/2 = -7.25 div. The emf difference is -7.25 on the 30-pV

range (60 div full scale). If the corresponding reading of -12 divisions on the 100-division scale is multiplied by 0.6 from table 2, essentially the same result is obtained.

611
The table gives factors, K, for converting deflection changes to mV, jjV , and

nV for the four most-used ranges. As indicated in the text, a conversion to

mV is preferred for results in percent while nV are more convenient for ppm.

3.8 Peak and Average Responding Instruments

Peak and average responding instruments are widely used for certain applications, and their principles can be adapted for ac-dc comparisons. They

are not electrothermic and are therefore outside the intended scope of this

writing. However, since at least one peak comparator [11], a peak calibration

method [12], and one average responding ac-dc comparator [13] have been

developed, they are noted here and referenced. The accuracy of these com- parators is sufficient for many applications; however, they are more limited

in voltage and frequency range than electrothermic instruments. These com- parators are not commercially available at this time.

4.0 Switches and Potentiometers

As noted earlier (3.0), some commercially available multirange TVCs have

provisions for ac-dc switching as well as an adjustable emf source and a detector for monitoring the thermoelement output. If these components are not

included, or if single-range TVCs, thermoelements or TCCs are to be monitored,

the equipment described in this section is not difficult to build, and the

parts are readily available. The circuits are shown in some detail in fig. 3,

and of course, variations can be made to meet particular needs. The diagram

of the potentiometer includes two optional features, as noted below the

caption, which were discussed in previous sections.

4.1 AC-DC Switches

The ac-dc switch shown in fig. 3 connects the TVC to either direction of dc

voltage, or to the ac voltage to be measured, in the order suggested in

section 3.2. Other switch arrangements, such as reversing dc with a separate

switch, should be satisfactory also. The ac input to the switch, and the

cable connection to the TVC, should be coaxial to minimize electromagnetic

interference and circuit impedance. The switch and the potentiometer should

be in separate shielded enclosures, and the cable to the detector, D, input

should also be shielded.

The dc voltage may be supplied by a calibrator which serves as both source

and standard, or a separate standard (a volt box and potentiometer or a digital voltmeter) may be connected at the terminals for "DC Std." The

standard should not be appreciably affected by changing the polarity of the

applied voltages. At very low voltages, where the voltage drop across the

line resistance may be significant, the dc voltage should be monitored at

the "DC Std." terminals. The cables with conductors c and shields s (fig. 3)

should have equal resistance (identical cables are preferable) to equalize the

voltage difference between the ac and dc voltage terminals and the com- parison points at the switch contacts.

The ac-dc circuit should have only one ground point, usually, but not

necessarily, at the ac source. The switching circuit shown in fig. 3 provides

612
a direct connection from the TE to ground. It is preferable to have no switch

in this part of the circuit. If a switch is used, and if it opens an instant

before the switch contact in the input side, the insulation between the

thermoelement heater and the thermocouple will probably be punctured by an excessive voltage.

4.2 Lindeck Potentiometers

A two-range (10- and 20-mV) potentiometer is shown in fig. 3. It is powered

by a 1.35-V mercury battery and controlled by two helical resistors, which may

be either the 3- or 10-turn devices currently available. If the "deflection

method" proposed in section 3.2 is used, the fine (10-ohm) control need not

have resolution sufficient for an exact null. The controls are adjusted for a detector indication in the portion of the scale where the first reading (e ls section 3.2) should be made.

If the 1-mA meter (M) is omitted, the instrument is a balancing emf source which cannot measure the TE output. However, as noted earlier (3.2), the emf

can be measured by the detector if the detector is a microvol tmeter. Select a range to measure 10 mV or more, turn the balancing emf off, and close the key

(ki).

As noted in the caption (fig. 3), the resistors R 4 and R 5 (section 3.3) and R 6 and R 7 (section 3.6), and their associated switches, are used to adjust and measure the sensitivity of the detector, D. These components may be omitted

if the detector is a microvol tmeter , or if a null method (3.1) is used.

The potentiometer and detector should be grounded, preferably at the negative

input to the detector. This is sometimes done by linking the detector input

to ground through the detector's power cord. A few instruments may have the

negative output connected to the instrument casing which is grounded. However, it is preferable to connect the circuits to the grounded potentiometer

shield by closing the switch at the negative input terminal (fig. 3), or by some other link at this point.

4.3 Detector Keys

The detector keys ki and k2 , and the three associated resistors (fig. 3), are typical of those used in these procedures. Key k 1 should have low-thermal

contacts and low contact resistance. A two-position lever switch is recom- mended for k2

. The contacts should be good, but a quality less than that of

k x should be satisfactory. The resistor R x is always in parallel with the

detector, unless k x is closed for a detector reading. This keeps the detector

"quieter" when the key is open. Rj also serves to reduce the detector sensitivity when k2 is closed for preliminary balance adjustments. It should not

be necessary to switch a multi range detector to a higher range for preliminary

balances with this key configuration. An Rj of 100 ohms is satisfactory for

most high-impedance detectors. R 2 and R3 should be chosen experimentally,

depending on the range most often used and individual preference. However,

20 kfi and 100 kfi, respectively, may be useful as values to start with.

613
5.0 AC-DC Difference

The ac-dc difference in a thermal voltage converter is nearly all due to

reactance in the ranging resistors, the range selector switch, and other

switches and connections in the ac-dc circuit. Series inductance will impede

the ac current, so that more ac than dc voltage is required for equal TE

current--and thermocouple output. The sign of the correction is, by eq. (4),

positive in this case. A positive correction can also result from ac current

being lost to ground by capacitance. However, capacitance between resistors

connected in series, or between turns of wire in a wire-wound resistor, will

increase the TE current for a given ac voltage and result in a negative

correction. Compensation for this effect is sometimes provided by adding

capacitors to ground which bypass the TE and part of the series resistors.

5.1 AC-DC Difference Corrections

AC-dc difference corrections can be plotted conveniently on a semilog scale as

in fig. 5, and such a plot can be used to find good estimates of corrections

at frequencies where determinations were not made. Fig. 5 illustrates a typical set of correction curves where the specified error limits are +a to

-a. Low-voltage ranges in a multirange TVC usually have small corrections

(curves A, B, and C). Intermediate and high-voltage ranges are often limited

to lower frequencies (curves D and E), because the effects of reactive

components are larger and more difficult to control. The useful frequency

range at high voltage is often extended by adding compensating elements as mentioned above. Optimum compensation (curve F) will sometimes result in a change in direction (and sign) of the correction, and the change in magnitude

is usually rapid after the sign change.

Figure 5. AC-dc difference correction curves.

614
It is usually sufficient to make only one determination of ac-dc difference at

low and intermediate-voltage ranges. This determination should be at the

highest frequency of interest. If the instrument is used only at audio

frequencies, a test at 20 kHz is often sufficient, even on high-voltage

ranges. The correction curve can be expected to approach zero rapidly, and if

the instrument is designed for use at 50 to 100 kHz, the correction should be

quite negligible at 10 kHz and lower. Where corrections are relatively large, or if there is evidence of a curve such as F (fig. 5), additional determinations should be made.

5.2 AC-DC Difference in Thermoelements

The low-range thermoelements used in TVCs (2.5 to 10 mA) rarely add significantly to the ac-dc difference of the instrument. The TE response to ac current is affected by two or three thermoelectric phenomena which, incidentally, are quite independent of frequency. However, the result of these

effects is less than 5 ppm for most low-current TEs. It should not be

necessary to redetermine the ac-dc difference of a TVC after a TE is

replaced, and new TEs furnished by the instrument manufacturer should not

require ac-dc difference tests before being installed. However, if they are tested, they should be treated as current converters (i.e., tested in series

with a standard TE), as they respond to the current through the series

resistors. The TE acts more as a voltage converter on very low ranges where

its impedance is a large part of the total.

AC-dc difference in higher-range vacuum TEs is larger due to skin effect in

the current conductors leading to the heater. The copper-coated nickel alloy

widely used to pass current conductors through glass seals is sufficiently

magnetic to cause a small increase in impedance. Therefore, more heating is

produced with ac current than with equal dc current. AC-dc difference resulting from skin effect usually varies from about 0.005 percent at 200 mA to

between 0.01 and 0.02 percent at 1 ampere—the highest range now available in vacuum bulbs. The effect in the low-current TEs used in TVCs is nearly always

negligible. However, it may account for very small positive corrections in some 0.5- and 1-volt ranges, and in the low-voltage TVCs (0.3 to 1 volt) used

with ac shunts for current measurements. The shunt and TVC combination is

ordinarily tested and used as a unit. However, the ac-dc difference of the

TVC can be determined and appropriate corrections applied (see section 8.0 and

reference [16]).

TEs for currents of 10 amperes and higher sometimes have a tubular heater, or

a heater with a C-shaped cross section made from a flat material. The heater

is mounted between heavy copper support blocks with terminals at either end.

The ac-dc difference of these elements can be very much dependent on the

location of other conductors, particularly the return conductor, which is

usually placed close to, and more or less parallel to, the TE heater. AC

current will distribute itself for minimum reactance, and may therefore be

concentrated close to, or away from, the thermocouple which is attached to the

side or top of the heater. This nonuniform current distribution can cause

ac-dc differences as large as 0.03 percent. The ac-dc difference of

these TEs can be made constant by mounting both terminals on one end of the

615
device and connecting a rigid copper conductor between one terminal and the

heater support at the other end. This conductor can also be placed, relative

to the heater, for minimum ac-dc difference.

Practically all low-frequency (below 60 Hz) ac-dc differences in thermoelectric transfer instruments are due to the thermoelement [9]. At most

frequencies the heater temperature, and therefore the TE output, are essen- tially constant. However, at low frequencies the heater is cooled slightly

between peaks of ac current by conduction through the heater supports and

the thermocouple wires. This effect is greater in high-current TEs and may be

detected up to about 60 Hz on some ampere-range TEs. However, in 2.5- to

10-mA TEs used in TVCs the effect usually occurs below 5 or 10 Hz. It also

decreases sharply as the heater current is reduced on all current ranges.

Correction curves for three current levels in one TE are illustrated in the

left-hand portion of fig. 5.

6.0 AC-DC Difference Measurements

AC-dc difference corrections to a transfer instrument (TVC or TCC) under

test, 6. , are determined relative to a similar standard whose corrections, 6 , are known. Corrections to a standard or reference TVC or TCC are determined

at the National Bureau of Standards and other laboratories. Measurements of

6.-6 may be made by either of the two methods outlined below. Fig. 6 shows a typical test arrangement for comparing two TVCs. AC and the two directions of

dc voltage are connected to the TVCs successively by the ac-dc switch. Note

that both TEs are connected directly to ground as in fig. 3. The switch shown

in fig. 3 can also be used in ac-dc difference measurements. However, the

switching sequence indicated in fig. 6 may be preferable, and the switch is a bit simpler. The TVC outputs are monitored with detectors (D) after being

balanced by the emf sources (B), each of which may be a potentiometer, as in

fig. 3.

DC

AC

Figure 6. Circuit for ac-dc difference measurements.
The power supplies need not be calibrated; however, if the "null method" is

used, provisions must be made to measure the dc voltage. The dc supply must

have fine controls, with resolution of a few ppm, for adjusting the input to

the test TVC for equal output on both directions of dc voltage. AC voltage

may be supplied by an oscillator, amplifier, and appropriate transformers.

Fine control for the ac voltage is necessary if the deflection method is used.

Control circuits are discussed in section 6.4.

6.1 A Null Method

AC-dc difference can be measured using a procedure in which only null

readings are made on the detectors used with the TVCs. This method is similar

to the "null method" described in section 3.1 for voltage measurements. In

this procedure, ac voltage is applied first, and both balancing emfs are adjusted to null the detectors. DC voltage is then applied, adjusted for a null on the test TVC detector, and measured (or read from a dc calibrator).

The dc voltage is then readjusted for a null on the standard TVC detector and

measured. In the example below, these voltages are listed on the line labeled

"DC+." DC voltage of the opposite polarity is then applied, adjusted for the

two nulls, and measured as before. ("DC-" in the example.) The ac voltage

should then be reapplied, and any drift from the original null position on either detector should be evaluated for significance. If either TVC output is

drifting appreciably, more warm-up time should be allowed, or the deflection

method (6.2) may be used.

The relative ac-dc difference between the two TVCs in volts is equal to the

difference between the average of the dc voltages required for nulls on the

standard TVC and those required for nulls on the test TVC. The difference is

usually converted to percent (or ppm) as in the following example.

Voltage Voltages required for detector nulls

Test TVC (V

t

) Std. TVC (V

g

)

DC+ 200.000 199.952

DC- 200.032 200.010

Ave DC 200.016 199.981

AV = V -V. -0.035 V

s t

Relative ac-dc difference -0.018%

6 (Std.) known +0.003%

6*(Test) -0.015%

t" or -150 ppm

This procedure requires no galvanometer calibration or measurements of n. However, the adjustments are tedious and time consuming, dc standards are necessary, and the disadvantages discussed earlier (3.1) are present here

al so.

6.2 A Deflection Method

The deflection method for making ac-dc difference measurements is similar to

the one described in section 3.2 for making ac-dc voltage comparisons. DC

617
voltage is applied to both TVCs, and the balancing emfs are adjusted to null

both detectors. AC and the two directions of dc voltage are adjusted in turn

to obtain a null indication on the detector used with the test TVC. The

corresponding indications on the standard TVC's detector are recorded, and an emf difference, E g

- E^, is computed as in section 3.2. The ac-dc difference

of the test TVC, 6 . , (defined in section 2.4) is determined by evaluating E

- t a E^/nE and applying a correction for the ac-dc difference of the standard TVC,

V

Vad- V . E ad- E . 6. = 6\ V

d

nE s "t

In this application, V

g

and V^ are the ac and dc voltages required for equal

response from the test TVC. The emf difference, E g

- E^, can be computed from

readings taken from a digital microvol tmeter or a deflecting detector, as discussed in section 3.2, and the scale reading techniques described in

section 3.7 are useful here also. The reading sequence suggested for voltage

measurements (DC+,AC,DC-) may be used for ac-dc difference measurements.

However, it is probably advantageous to take four readings in the sequence

suggested in fig. 6 (AC,DC+,DC- ,AC). Either sequence will minimize the

effects of drift in the TVC outputs, if the readings are taken at time intervals that are about equal. The second reading with ac voltage should increase

the reproducibility.

A measurement made with a deflecting detector can be illustrated using the

indications from fig. 4. If it is assumed that there is no appreciable drift,

so that both readings on ac voltage are equal (i.e., 36 divisions), then

AD = (D

2

- D

1

- D

3 + D2 )/2 = (36-23-73+36)/2 = -12 div.

If nE = 16.1 mV, 6^ = +0.003% and K = 2 x

10" 4 mv/div (10 uV rangetable 2), then s

6

t = -0.015 + 0.003 = -0.012%.

6. 3 TE Comparators

Stable power supplies are necessary for accurate measurements, and instability

is probably the major cause of nonreproducibi 1 i ty in ac-dc difference com- parisons. This difficulty can be largely overcome by using a TE (thermoelement) comparator. The instrument is not commercially available at this

time; however, it is not difficult to build. A brief description of the

general principle and advantages is given below. A more complete discussion

is left to the referenced papers. TE comparators have been built in at least

three configurations [9,10,14], but they all employ a divider circuit to which

the emfs from the test and standard TVCs are connected. The divider is

adjusted, nulling a detector, and at this point the divider setting corresponds to the ratio of the emfs.

618
The comparator sketched in fig. 7 shows a typical circuit. The potentiometer

(Pot'r) is used, with key k x , to measure both emfs and to monitor the test TVC

output. The divider (DIV) is a two-dial Kel vin-Varley , the second dial being

a ten-turn helical resistor. The lower emf, which may be either from the test

or standard TVC, is balanced against a portion of the higher emf by adjusting

the divider. A balance is indicated by a null reading on the microvol tmeter

(uVm) with key k2 closed. After this preliminary balance, the ac and dc

voltages are applied in the sequence indicated, and each voltage is adjusted

to give a test TVC output equal to the Pot'r emf. Then with key k t open, k2 is closed and an emf is read from the uVm. These emfs differ because of ac-dc

difference and dc reversal difference. If both directions of dc voltage are

used, the ac-dc difference can be computed from them.

Figure 7. TE comparator with two TVCs and ac-dc switch.

Small fluctuations in the power supply output will produce nearly equal

proportional changes in the emfs, and the divider balance will not be affected

appreciably. The stabilizing effect is greater if the TEs have well -matched

time constants and response characteristics (n values), but even if they are not quite matched, the effect usually affords a significant advantage. A

second advantage is that the ac and dc voltages do not have to be adjusted

for equal output from the test TVC so exactly as in other test methods.

Actually, if the test and standard TEs were quite similar, and the dc

reversal differences were small, it should not be necessary to adjust or monitor the test TVC output at all, once the ac and dc voltages are adjusted to be nearly equal. However, TEs usually differ enough to make

adjustments to near equal output necessary for accurate measurements of

ac-dc difference.

An automated model of this comparator is described in reference [15].

6.4 Voltage Control Circuits

Measurements of ac-dc difference require very fine voltage adjustments to

produce equal output from the test TVC. Fine controls are required for dc

619
voltage in the null method, and for both ac and dc in the deflection

method. AC and dc voltages are often supplied by five- and six-dial

calibrators. The lower two or three dials can be adjusted for a detector

null; however, this method is not very satisfactory. The knobs are often

small, close together and not easily turned. An exact null may not be obtainable with the resolution available, and it is sometimes difficult to place the

supply within easy reach of the operator.

The series resistance control shown in fig. 8 is recommended as a substitute

fine control for voltages from 1 to 1000 V. It is placed between the dc

supply and the ac-dc switch for dc voltage control, or between this

switch and the TVCs to control both ac and dc voltage. The voltages are

set to be nearly equal with the regular controls, and the final fine adjustments are made with this control. It can be built into a shielded box and

connected into the line by a two-conductor shielded cable. It can then be

placed where most convenient for the operator. It is important to remember

that the control carries the test voltage which is often high enough to be

dangerous. The casing must therefore be well insulated, and the shield firmly

grounded. This control can, of course, be used with other power supplies

which do not have sufficient fine controls. The voltage across the control,

and hence the voltage at the TVC terminal, is adjusted with the ten-turn,

1-kft, helical resistor R t

. The sensitivity is adjusted by shunting R 1 with

any one of 10 resistance values R2. The value of R2 is selected by a rotary

switch, and is increased* as the voltage increases. A resistor R3 of about 250

ohms prevents R 2 from being shorted and improves the linearity. The most

appropriate setting for R2 depends not only on the voltage level and user preference, but on the current drawn by the TVCs —usually 5 to 20 mA. The

recommended value of resistance for each switch setting is listed in table 3

below. Each successive step increases R2 by 50 to 70 percent.

250 Ci Supply

2

1000 Q,

to TVCs

15ft

Figure 8. Series resistance fine control for ac and dc voltages.

620
Table 3. (see fig. 8)

Switch Resistance of R« Switch Resistance of R 0

position (ohms) position (ohms)

1 15 6 150

2 25 7 250

3 40 8 400

4 60 9 650

5 100 10 1000

* * * * *

AC voltage and current from an oscillator-amplifier supply can be controlled

with the circuit shown in fig. 9, which regulates the input signal to the

amplifier. Resistor R x is a ten-turn helical resistor for coarse control, and

adjusts the amplifier input from zero to near the oscillator output voltage.

The control is less coarse if this resistor is operated near full range. The

oscillator gain should therefore be set for an amplifier output only slightly

higher than that necessary for the desired output voltage. An R 1 of 1 kfi is

suitable for most oscillators, but 10 kfi may be necessary for those with

relatively high output impedance.

Oscillator

to Amplifier

Figure 9. Control for an amplifier input.

Fine control is provided by a ten-turn helical resistor R2 of 100 ohms and a shunt R3 which regulates the sensitivity. The value of R3 depends on several

factors in addition to personal preference, but 20 to 50 ohms is usually

satisfactory. The control assembly should be well shielded. If it is connected

to the oscillator and amplifier through a shielded three-conductor cable, it can be moved around and placed conveniently for the operator.

621
6.5 Verifying AC-DC Difference Corrections

AC-dc difference corrections are primarily due to reactance in the switches

and resistors in the TE circuit (see section 5.0). They are not likely to

change, if these components are rigidly mounted and spaced so that capacitance

between components, and between components and ground, is not excessive.

Changes are probably more likely, if fixed and adjustable capacitors are added

for frequency compensation. This is often done in multi range TVCs for the

higher voltage ranges. If repairs or alterations are made that might affect

these circuits, the ac-dc difference corrections should be redetermined.

If there is no reason to suspect that a change has occurred, then

recal ibration at NBS or other laboratories can be several years apart.

(Five-year intervals are reasonable.) However, it is advisable to make

periodic comparisons between TVCs within a laboratory to detect any changes

that might have occurred. Such tests are especially valuable at high voltages

and frequencies where changes are most likely.

Tests are made most easily between adjacent ranges in a set of single-range

TVCs, or between members of two sets. If these tests are made accurately, it

should not be necessary to send the entire set to another laboratory for

recal ibration. A few ranges, perhaps a low, medium, and high range, would

serve as a basis for determining corrections to the other ranges by intercomparison. Of course, other sets and multirange instruments can be compared

with such a set.

If two multirange TVCs are available, one can be calibrated relative to the

other. Such periodic comparisons will assure that no changes have occurred as long as the results are the same from one test to the next. It is very

unlikely that both instruments will change equally and simultaneously.

If only one multirange TVC is available, no intercomparisons can be made.

However, the procedure outlined below should be useful in detecting any

significant changes in the ac-dc difference corrections. It consists of

measuring the difference between a constant ac voltage and the average of the

two polarities of a constant dc voltage at rated voltage on a given range and

then immediately with the same voltages on the next higher range (e.g., 50

volts on the 50-volt and the 100-volt ranges). If the voltages are stable

during the test, the difference between the ac-dc difference corrections of

the two ranges (& 1 - 6 2 ) can be computed. The voltage difference can be

determined by evaluating (E - E d

)/nE as in section 3.2, and from the equation

in section 6.2,

+ 6.

V

d

nE

Then, since (V - V,)/V, is the same,

E al

- E dl

1

E 1

where the subscripts 1 and 2 refer to the two ranges.

622
This procedure is more likely to be used on instruments with built-in detectors and balancing emf circuits, so that E and n values may not be known.

However, the detector can be calibrated directly by making a small change in

the dc voltage, observing the resulting deflection change, and computing a multiplying factor, K, (ppm/div) as suggested in section 3.6, first paragraph.

In the example of actual data below, multiplying factors K x and K2 were

determined for the 50- and 100- volt ranges of a typical multi range TVC. The

sensitivity of the 100-volt range was, of course, low (38 ppm/div) as only

50 percent of rated voltage was applied. AC voltage at 100 kHz and the

two polarities of dc voltage were then applied in the sequence

DC+,AC,DC-,AC,DC+,AC,DC-, and seven deflections, Dj to D 7 , were observed.

Then

D a = D

2

- (Oj+Dj)^, D

b = D4

- (D

3

+D

5 )/2, Dc = Dg - (D5

+D

? )/2 ,

and "ave AD" is the average of the three. K is in ppm/div and

6

1

- 6

2 = (AD

2 x K2

) - (AD

1 x K^) ,

where the subscripts 1 and 2 refer to the 100- and 50- volt ranges,

respectively.

Range ^1 ^2 ^3 ^4 ^5 ^6 ^7 ave AD K AD x K 100 23.0 14.0 25.0 14.0 23.0 13.5 25.0 -10.2 38 -390

50 27.0 12.5 23.5 12.0 25.0 12.0 22.5 -12.2 8.6 -100

The ac-dc differences of the 100- and 50- volt ranges at 100 kHz were +210 and

-90 ppm, respectively, when measured by conventional methods. These corrections differ by 300 ppm, and the voltage comparisons indicate a difference (6 t 62 ) of 290 ppm. This agreement is closer than in most of these tests; however,

changes on the order of 50 ppm should be detected readily enough by this

method.

Accurate knowledge of the ac and dc voltages is not necessary, and the

need for equality is only that all readings be within the range of the readout detector. It is necessary, of course, to ensure that neither voltage

changes significantly during the few minutes required for a measurement.

Changes are more likely when the TVC range, and hence the load on the

supplies, is changed. However, the loads are very small —usually 2.5 and 5

mA—and most ac and dc calibrators are well enough regulated to remain

stable during these changes. An additional load resistor can be inserted in

parallel with the TVC for a few seconds to roughly double the current drawn.

No significant change in the TVC detector balance should be observed. A

repeat test after a few minutes with consistent results will indicate that one voltage is not drifting relative to the other.

If stable calibrators are not available, the voltages may be monitored with

digital voltmeters. However, most ac DVMs currently available are limited

to 4-1/2 digits, and this resolution is marginal at best in this application.

623
7.0 Measurements of n

The measurements described thus far are made to determine small voltage and

current differences by evaluating (E -E ,)/nE. As noted earlier (2.1), the

o Q

factor n is near 2 at very low currents, but usually 1.6 to 1.9 at rated

currents. The expression above can be evaluated if n is predetermined in a special test using dc voltage or current. Values of n corresponding to any

value of E can be found, if a curve is plotted of n versus E. Four or five

determinations between 50 and 110 percent of rated heater current are sufficient.

From eqs. (2) and (3),

n= EWI) .andforaTVC, n=

E (AV/V)

Measurements of n are made by introducing small measured proportional changes

in TE heater current (Al/I) and measuring the resulting output, E, and changes

in output, AE. Current changes should be large enough for good resolution,

but small enough to keep all detector readings on scale. Changes of a few

tenths to about one percent are usually convenient.

These tests are probably made more conveniently on the 100-V range of a multi range TVC, and n values obtained on one range can be used with any range,

if they are correctly related to the TE output. Low-voltage, single-range

TVCs can be placed in series with additional resistance, and more manageable

voltages can then be applied. Low-range TEs (2.5 to 10 mA) may be connected

in series with 50 to 200 kQ and appropriate voltage applied for the desired

heater currents. The voltages may be supplied, and small voltage changes

"dialed in," from a dc calibrator. Or the voltage may be supplied by a stable dc power supply with fine controls, and small changes measured with a digital voltmeter or a volt box and potentiometer.

In the example below, one determination of n is made by applying 90.000 and

90.500 volts for a proportional change of 0.00556, and E = 8.51 mV. Imbalance

emfs, e, are read from a digital microvol tmeter , and three emf changes are computed from seven readings.

AE^- (e-^ep^, AE

2=e

4"(e3+e5 )/2 , AE

3=e6

-(e

5+e

? )/2

Successive readings are made at the alternate vol tages--90. 0 and 90.5. The

average of the three emf changes (80.4 pV) is converted to 0.0804 mV for the

calculations.

V AV E 6

1

6

2

6

3

6

4

e

5 f7_ ave AE AV/V n 90 0.5 8.51 12.3 92.8 11.4 91.2 10.8 90.8 10.4 0.0804 0.00556 1.70

Current changes can also be introduced in a TVC, or in a TE in series with a high resistance, by inserting a known resistance change in the TE circuit. A

relatively low resistance, with a switch for shorting it, is placed in series

with a TVC or TE, and as the switch is opened and closed, AI/I=-AR/R. The

power supply must, of course, be well regulated, and the resistance ratio AR/R

624
should be known to about one percent. The calculations are simplified, as AI/I is the same at all voltage levels. The change can, of course, be a round

number. A 400-Q resistor in series with a 40- kQ ,5-mA, 200- V TVC will

introduce a change of 0.01 (one percent) at all voltage levels.

Currents higher than 10 mA are usually measured with a potentiometer and a shunt in series with the TE.

The TE output emf, E, can be measured with a Lindeck potentiometer (fig. 3), or

it can be measured by the detector, if a microvol tmeter with a 10-uV range is

used, by turning the potentiometer off and closing key kx

. Emf changes, AE, are measured by the detector as described in section 3.2.

It is difficult to measure n with an accuracy better than about one percent.

However, that accuracy is sufficient, as errors in n have a second-order

effect on the end result. A one percent error will result in an equal error

in measuring a difference, say of 0.05 percent, between an ac voltage and a dc reference voltage. However, this will affect the ac voltage measurement by

only 0.0005 percent (5 ppm).

8.0 Current Measurements

Thermoelements are sometimes used as thermal current converters (TCCs), and

they are usually excellent up to about 100 mA. They have ac-dc differences at

higher currents, as discussed in section 5.2, which can be evaluated, and

corrections can be applied. However, TCCs usually consist of a set of special

ac shunts and a low-voltage (0.3 to 1 volt) TVC. In some commercial models,

the shunt output is connected to the TE in a multi range TVC through a special

connector. These TVCs include a balancing emf source and a detector for

monitoring the output of the TE. Instructions for using these instruments

are, of course, furnished by the manufacturers.

Shunts can also be connected to a low-voltage, single-range TVC, and its

output can be monitored with an emf source or a potentiometer like the one shown in fig. 3. The detector may be any one of the four instruments discussed in 3.2, 3.3, and 3.4. Techniques for galvanometer calibration (3.6)

and detector readings (3.7) are also useful with TCCs.

AC current may be measured, and ammeters calibrated, with the equipment and

circuit shown in fig. 10. The procedure is very much like the deflection

method described in section 3.2 for voltage measurements. DC current is

adjusted to be nominally equal to the ac current, and the balancing emf or potentiometer (P) is adjusted for a near null indication on the detector (D).

The indication, e lf of the detector is recorded. The TCC is then switched to

ac current, the ac supply is adjusted for the desired indication on the

instrument under test (Test AM--fig. 10), and the detector indication, e2 , is

recorded. Reversed-dc current is then applied to the TCC, adjusted to the

nominal value, and e 3 is recorded. The difference between the ac and dc

currents is then computed from the equation,
where 6

g is the ac-dc difference correction to the TCC, and

E a

" Ed = e

2

" (e

l + e

3 )/2 '

If the detector is a deflection instrument,

E a

" E d = (D

2

• (D

1 + °3 )/2) x K '

DC Supply

and

Standards

AC

Supply

Figure 10. Circuit for current measurements

converter (TCC).

using a thermal current

The calculations illustrated in 3.5 are then practically the same as those for

current measurements. I is substituted for V, and TCC for TVC, in table 1.

The ac-dc difference of a TCC can be determined, relative to a standard

instrument, with the circuit shown in fig. 11. The equipment and switch may

be the same as those in fig. 10, but no dc standards are required. The

switching sequence may be as shown (DC+, AC, DC-), although the sequence

suggested in 6.2 (AC, DC+, DC-, AC) may yield greater reproducibility. The

switch shown in fig. 6 may be used if the current-carrying capacity is

sufficient.

Each TCC output is monitored with a balancing emf, or potentiometer (P), and

detector (D). A method similar to that suggested in 6.2 may be used. The ac and dc currents are adjusted for equal response (output emf) from the test

TCC, and imbalance emfs are read from the detector used with the standard TCC.

If four readings are taken, e x and e4 are read with ac current, and e2 and e 3

with the two directions of dc current. Then

626
E d = < e

l

- e

3 + e4

)/2,

E a

- E d = ((D

1

- D

2

- D

3 + D4 )/2) x K,

and

a d _ a d . ~ .

7 - p— + o — o

.

I . nE s t

d

where 6^ and 6

g are the ac-dc difference corrections to the test and standard

TCCs.

If the shunts have shields or casings (dashed lines in fig. 11), it is recom- mended that they be connected as shown in fig. 11. If one shield were attached between the shunts, a small capacitive current might flow between

that shield and ground. This current would flow through one shunt, but not

the other, causing an error.<5h

Figure 11. Circuit for ac-dc difference tests of thermal

current converters (TCC).

Additional suggestions for building and using TCCs are found in reference [16]

where the following are discussed:

(A) Members of a set of shunts can be intercompared very much as TVCs are (see

sections 6.0 to 6.5), and corrections determined for each shunt and TE com- bination relative to two or three ranges which have known corrections.

(B) A "short set" of shunts can be built in which each shunt in the set is

used with two TVCs (0.3 and 0.6 V) to form two current ranges. Thus the two

627
TVCs and six shunts provide 14 ranges. It is, of course, necessary in this

arrangement that the TVC correction be negligible or that a correction, 6y, be

applied. AC-dc difference in low- voltage TVCs is mainly due to skin effect in

the input leads to the thermoelement heater (5.2). This effect can be

practically eliminated by using TEs with platinum lead-in wires. Such

elements are available from one or two manufacturers upon special request, and

at a reasonable price.

(C) If the TVC corrections are appreciable, they can be determined relative to

other TVCs with known corrections. Then, at high and middle ranges, the ac-dc

difference of the shunt and TVC combination, 6p, is practically equal to the

sum of the ac-dc differences due to the admittance of the shunt, 6^, and the

impedance of the TVC, 6j. As discussed in the referenced paper, this

separation of correction components is not feasible at low currents (less than

0.25 A in most cases) where the TVC carries a substantial portion of the total

current. These low-range TCCs should be tested and used as shunt-TVC

combinations.

628
References

[I] "AC-DC transfer instruments and converters," American National

Standard C100.4, ANSI, 1430 Broadway, New York, NY 10018, 1973.

[2] F. J. Wilkins, T. A. Deacon, and R. S. Becker, "Multi junction thermal

convertor and accurate ac-dc transfer instrument," Proc. Inst. Elec.

Eng., Vol. 112, No. 4, pp. 794-805, 1965.

[3] F. J. Wilkins, "Theoretical analysis of the ac-dc transfer difference

of the NPL multi junction thermal convertor over the frequency range dc

to 100 kHz," IEEE Trans. Instrum. Meas. , Vol. IM-21, pp. 334-340, 1972.

[4] F. L. Hermach and D. R. Flach, "An investigation of multi junction thermal

converters," IEEE Trans. Instrum. Meas., Vol. IM-25, pp. 524-528, 1976.

[5] Manfred Klonz and F. J. Wilkins, "Multi junction thermal converter with

adjustable output voltage/current characteristics," IEEE Trans. Instrum.

Meas., Vol. IM-29, pp. 409-412, 1980.

[6] F. L. Hermach, "Thermal converters as ac-dc transfer standards for

current and. voltage measurements at audio frequencies," J. Res. Nat. Bur.

Stand., Vol. 48, pp. 121-138, 1952.

[7] F. C. Widdis, "Theory of Peltier- and Thomson-effect errors in thermal

ac-dc transfer devices," Proc. Inst. Elec. Eng., Vol. 109, Pt. C

(Monograph 497 M) , pp. 328-334, 1962.

[8] F. L. Hermach and E. S. Williams, "Thermal voltage converters for

accurate voltage measurements to 30 megacycles per second," Trans. AIEE

(Communication and Electronics), Vol. 79, Pt. I, pp. 200-206, 1960.

[9] F. L. Hermach and E. S. Williams, "Thermal converter for audio-frequency

voltage measurements of high accuracy," IEEE Trans. Instrum. Meas., Vol.

IM-15, pp. 260-268, 1966.

[10] E. S. Williams, "Thermal voltage converters and comparator for very

accurate ac voltage measurements," J. Res. Nat. Bur. Stand., Vol. 75C,

pp. 145-154, 1971.

[II] P. Richman, "A new peak-to-dc comparator for audio frequencies," IRE

Trans. Instrum., Vol. 1-11, pp. 115-122, 1962.

[12] D. R. Flach and L. A. Marzetta, "Calibration of peak ac to dc

comparators," ISA Conference Report No. 14.2-3-65, 1965.

[13] L. A. Marzetta and D. R. Flach, "Design features of a precision ac-dc

converter," J. Res. Nat. Bur. Stand., Vol. 73C, pp. 47-55, 1969.

[14] R. S. Turgel , "A comparator for thermal ac-dc transfer standards,"

ISA Trans., Vol. 6, No. 4, pp. 286-292, 1967.
[15] E. S. Williams, "A thermoelement comparator for automatic ac-dc

difference measurements," IEEE Trans. Instrum. Meas. , Vol. IM-29, pp.

405-409, 1980.

[16] E. S. Williams, "Thermal current converters for accurate ac current

measurement," IEEE Trans. Instrum. Meas., Vol. IM-25, pp. 519-523, 1976.

Other papers not specifically cited:

F. L. Hermach, J. E. Griffin, and E. S. Williams, "A system for accurate

direct and alternating voltage measurements," IEEE Trans. Instrum. Meas.,

Vol. IM-14, pp. 215-224, 1965.

F. L. Hermack, "AC-dc comparators for audio-frequency current and

voltage measurements of high accuracy," IEEE Trans. Instrum. Meas.

,

Vol. IM-25, pp. 489-494, 1976.

*U.S. GOVERNMENT PRINTING OFFICE: 1982 360-997/2265 1-3 630
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT. VOL IM-29. NO 4. DECEMBER 1980

A Thermoelement Comparator for Automatic

AC-DC Difference Measurements

EARL S. WILLIAMS

Abstract—AC-DC differences in thermal voltage converters (TVCs) are determined relative to a similar instrument, from imbalance F.MFs measured in a divider circuit to which both instrument outputs are connected. These EMPs are more stable than the input voltages

to the converters, because input voltage fluctuations produce nearly equal proportional changes in the converter outputs. The divider bal- ance is therefore affected only slightly. The divider and a Lindeck

potentiometer for monitoring the test instrument output are driven by

stepping motors and balanced automatically. AC and DC voltages are

applied by programmable power supplies whose outputs are adjusted,

in response to imbalance EMPs in the potentiometer circuit, to pro- duce equal output from the test instrument. The system automatically

calibrates one voltage range at one or more frequencies, but requires an operator to change ranges and to enter the test parameter at a computer terminal.

Introduction

AC VOLTAGE IS measured most accurately at the

. present time when it is compared directly with a dc

reference voltage using a thermal voltage converter (TVC).

These instruments, which are also called thermal transfer voltmeters, consist essentially of ranging resistors and a ther- moelement (TE), a short straight electrical heater whose

temperature is sensed by a thermocouple or a thermopile

Multirange TVCs usually include a balancing EMF source and a null detector for monitoring the TE output. The reference voltage Vd is usually an accurately measured

"round number" value which is nominally equal to the ac

voltage Va to be measured. The proportional difference be- tween the two voltages ( Va — Vd )/Vd is measured by observing

Manuscript received June 28, 1980.

E. S. Williams is with the National Bureau of Standards, Washington, DC

20234.

the output E of the TVC, and changes in output as the TVC

is connected alternately to the two voltages. Both polarities of dc voltage are generally used because the TE in most TVCs

has a significant reverse dc difference. When the difference between the ac and dc voltages is small

(Va ~ Vd) (Eg - Ed)

Vd nE

(1)

Ea is the EMF observed when ac voltage is applied to the TVC, Ed is the average of EM F's observed when the forward and reverse dc voltage is applied, and the factor n relates small changes in TE heater current to the corresponding changes in output EMF.

Ideally the TE has a square-law response with E = kl n (n = 2 and k constant for small changes in /). However, the value of n deviates from 2 as the heater current is increased and is usually in the range of 1 .7 to 1 .9 at the rated current of the TE.

Values of n must be determined at several levels by applying

known voltage changes AV/V to the TVC and observing the output E and changes in output AE. The equation is very much

like (1 ) above: AV/V = AE/nE

or

n = AE/E

AV/V

(2)

(3) AC voltage measurements, as described above, usually re- quire the application of a correction (<5) for the ac-dc differ- ence of the TVC. Such corrections are needed due to induc- tance in the ranging resistors and range selecting switches and

631
IEEE TRANSACTIONS ON INSTRUME

Fig. 1 . Automatic test system with comparator in the foreground.

to capacitance between these components and ground. The TE

rarely contributes significantly to the ac-dc difference except

at very low frequencies. AC-DC difference corrections of a TVC under test b, are determined by comparison with a standard TVC whose corrections hs are known. Measurements

of 6, — 5, are made by applying the same ac and dc voltages

to the two TVC's, adjusting the voltages for equal output from

the test instrument, and observing the changes in response of the standard. The output ot the test TVC is usually balanced by an adjustable EMF source and monitored by a detector which may be a microvoltmeter or a calibrated galvanometer.TE (thermoelement) comparators have been developed at NBS [1], [2] and elsewhere which have certain advantages

over other methods for making these measurements. The

system described here is built around a new automatic comparator which utilizes a digital millivoltmeter to measure TVC

outputs EMF and a digital linear amplifier (DLA) to measure

imbalance EMF's as small as 10 nV. Both instruments have BCD outputs. The system is interfaced with a minicomputer

by a Modular Interactive Data Acquisition System (MIDAS)

which was developed at NBS [3]. Similar programming for another interface scheme, such as the IEEE 488 standard bus

for instrument control [4], should present no particular

problem.

Fig. 1 is a photograph of the console showing the comparator, on the right table top, connected to two TVC's. Equip- ment in the rack are, from top to bottom, the MIDAS inter- face, a panel containing the millivoltmeter and certain relays, the DLA and the ac-dc switching circuits. Not shown in the photograph are the power supplies, a cathode ray computer

terminal (CRT), and a teletype (TTY) for printing test results.

TE Comparator Principle

The automatic comparator is similar to the NBS manual

model described in reference [2]. Both comparators consist

essentially of a 1000-fl divider and a Lindeck potentiometer.

In the manual comparator (Fig. 2), the higher EMF is con-6

TATION AND MEASUREMENT, VOL. IM-29, NO. 4, DECEMBER 1980

AC DC* 0C- AC

o o o o

Fig. 2. Manual TE comparator circuit with TVC's.

nected to the two-dial Kelvin-Varley divider through the Eh

input and the lower through El. The potentiometer may be connected to either input by switch S, and both EMF's are measured by adjusting the potentiometer for a zero reading on the microvoltmeter (/uV • m) with key k\ closed. The po- tentiometer is then connected to the test TVC output, which

may be either lower or higher than the standard output. The

divider is adjusted to balance the lower EMF, as indicated by

a zero reading on the ^<V • m with key k 2 closed. Then, as ac and dc voltages are applied to the TVC's and adjusted for equal

output from the test instrument (zero reading on the jtV • m

with k] closed), imbalance voltages in the low EMF circuit are measured by the yuV • m, and

8i _ ds =

iEA^E£ ) (4) ns EL Ea and Ed are the readings on the juV • m obtained with ac

voltage and the average of the readings with two polarities of dc voltage, EL is the lower EMF, and ns is the n characteristic of the standard TVC at its output EMF. Switch 5 has two

additional poles which reverse the connection to the /uV • m.

This is necessary to maintain a consistent sign convention for the EMF difference Ed — Ea [2]. Imbalance voltages in the divider circuit (key kj closed) are

not affected appreciably by changes or fluctuation in the input

voltage to the TVC's. Such changes will produce nearly equal

proportional changes in the two EMF's, and the divider bal- ance is affected only slightly. This stabilizing effect accounts

for the two advantages of the comparator: The input voltage

instability is less troublesome, and the requirement that the

voltages be adjusted for equal output from the test TVC is less severe. If the response characteristics of the TE's in the TVC's

are identical, or nearly so, equal output from the test converter may not be necessary. Most elements are similar enough to give the comparator a considerable advantage. However, if both

the reverse dc difference and the n characteristic differ at the same time, as they often do, a significant error in the ac-dc

difference determination can be expected unless the voltages

are adjusted for equal output from the test TVC.

Automatic Comparator

The divider in the automatic comparator is connected to either input, TEST or STD, by switch S (Fig. 2), which is turned by a 30° stepping motor, SMI. A 3'/2 digit millivolt- meter (mV • m) is connected across the divider by closing the

2-pole relay k\, and it measured the standard TVC output Es 2
WILLIAMS: THERMOELEMENT COMPARATOR

j*c supply [-

t

-

r^n

-

I TTY |

[ RELAY SWITCHING |—

[

MIDAS Inttrtace 7S 7\ r

(SM2j (SM3J

1-

Fig. 3. Diagram of automatic system with comparator.

in switch position A and the test TVC output E, in position B. The denominator in (4) (ns Ei) is computed using the lower EMF, and ns is computed from a equation which relates the values of ns to the output of the standard TVC. The divider is then connected to the higher EMF. The position of switch S

is determined by the MIDAS which detects a ground in the "/I" position of the switch. In this way, the EMF's Es and E,

are distinguished from one another, and the sign of the EMF

difference Ed - Ea (4) is also determined.

The 1 5-mV potentiometer is used only to monitor the output

of the test TVC, and it is therefore permanently connected to the TEST input, as shown in heavy lines in Fig. 3. The potentiometer current is supplied by a mercury battery and adjusted

by a ten-turn helical 3-terminal resistor for a near balance with one direction of dc voltage applied to the TVC's. The control

resistor is driven by stepping motor SM3 (Fig. 3) with 200

steps per revolution. The resolution is not high (about 8 ^V);

however, an exact balance is not necessary because the ac and

dc voltages are adjusted during the measurement procedure

for a balance between the test TVC output and the potenti- ometer. The steps necessary for a near balance are computed

from voltages measured by the mV • m with relay ki closed. Below 20 fiV, additional adjustments are computed for a balance within ±4 ^V from measurements made by the DLA.

The DLA is connected into the potentiometer circuit for these measurements by relay /c 4 , a low-noise key-type switch which

is closed in either of two directions by small solenoids. A two-dial Kelvin-Varley divider (Fig. 2) is used in the manual comparator to provide sufficient fine adjustment to be compatible with the narrow range and linearity limitations of the microvoltmeters or galvanometers used with that system. The wider (40-jzV) range of the DLA permits the use of a

single ten-turn resistor as a divider in the automated system.

The resistor must of course have a low-thermal low-noise moving contact. However, as no movements are made during

the actual measurements, the resistors currently available have

been adequate. The divider is balanced by stepping motor SM2

in procedures similar to the one used with the potentiometer.

The required steps for a near null are computed from imbal- ance voltages measured by the mV • m with relay fc 3 closed.

1 kO

1 kf)

Fig 4. AC-DC switching circuit.

When the imbalance voltage is sufficiently small, the DLA is connected into the divider circuit by relay k 4 , and additional adjustment are computed from its measurements of the re- sidual imbalance voltage.

Voltage Switching and Control

AC and DC supply voltages are connected to the TVC's by

the circuit shown in Fig. 4. Up to 100 V are supplied by the AC-L terminal by a programmable ac calibrator, and a companion high-voltage amplifier provides voltage at AC-H from

100 to 1000 V. The input to be used is selected manually by

switch S i

. The TVC's are connected to the AC-DC 2 terminal,

where the circuit can be opened by switch S2, and a resistance of 1000 or 2000 can also be added to the circuit. The resis- tance permits the power supplies to run in the range of 10 to 40 V where they are more stable and controlled more readily, while supplying less than 10 V to the TVC's. The terminals DC-Std, AC, and AC-DC 1 are used for voltage measurements

with one TVC. Relay k \ reverses the dc voltage to the TVC's,

and k 2 connects them to ac or dc voltage. These relays are shown in the unenergized position with dc voltage applied to the TVC's. The relays are activated by 26-V dc which is con- nected to the relay coils by programmable switches in the MIDAS controller. The power supplys are remotely programmed and turned on in response to the test voltage and the frequency (in kHz)

which are entered at the CRT along with other test conditions. The alpha-numeric "string" codes which activate the power

supplies are either stored in the computer or computed by it. Zeroes are added to the value of test voltage to extend it to seven digits for the dc supply which is a seven-dial calibrator and to five digits for the five-dial ac calibrator. Both calibrators have several ranges but only the 10-, 100-, and 1000-V ranges

are used. The codes for range selection are found by executing

three Basic language statements which relate the code to the

desired voltage range. The ac calibrator supplies frequencies from 10 Hz to 100 kHz in four decade ranges with eleven settings (0 to 10) in each

range. Thus there are 44 discrete frequencies available, and

each one is designated by a two-digit code. The code numbers

are computed from one of four equations which is selected according to the required frequency range. When the "round number" ac and dc voltages are applied

to the TVC's, both voltages usually require small adjustments

to make the test TVC's output equal to the potentiometer

EMF. The potentiometer setting is established as outlined

previously, using one direction of dc voltage as input to the 633
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-29, NO. 4, DECEMBER 1980

* DC im> -its.4 •>8«S25 -160.4 -»6«8MI «•*

-°8987 -213.2

- - 175 - 161 43 - 213 1 108

-98987 -213.2 -9899555 -176 -9898771 27.Z -98997 -229.4

- - 213 - 176 27 - 229 2 117

-98987 -229.4 -9899664 -190.4 -9898696 0 -98937 -242.8

- - 229 - 190 0

- 243 3 112

10 - 0 108 117 112 4 112

-mTs.TtsT.TE.iwa.nio i,nio 2 -'200, 200, n, 200, 100, 50, Fig. 5. Calibration data as displayed by the computer terminal.

TVC's. The test TVC often has a significant reverse-dc dif- ference which then requires readjustment. The ac voltage may

need adjustment because of the ac-dc difference and also be- cause the ac supply voltage may not be equal to the voltage at the TVC terminals due to reactance in the conductors, in the

relay k 2 (Fig- 4), and in the switches and S 2

- The voltages

are adjusted in response to EMF E\ measured by the DLA

while it is connected into the potentiometer circuit by relay k 4 (Fig. 3). The adjustment A V is computed using an adaptation

of (2) which may be written

AV = -(AE/nE)V. (5)

A nominal value of 1 .8 is substituted for n, E is the EMF of the

test TVC, and the negative sign gives the voltage change the proper direction. The numerical value of the voltage change

is calculated to give an EMF change AE equal to E\ and thus reduce E\ to near zero. It is then suitably formatted by the

controller to reset the five-digit ac supply or the seven-digit dc

supply. Unless the initial change is large, a single adjustment

is usually sufficient for a balance within 1 /jV.

Fig. 5 is a photograph of the CRT display of actual test results. The data appears on the screen one line at a time as the

test proceeds. The left-hand column shows the adjusted value

of voltage (five digits for the ac supply and seven for the dc

supply), and opposite each voltage is the EMF measured by

the DLA in the divider circuit. The first four EMF readings

are repeated on the fifth line, and the first value of ac-dc dif- ference 108 is listed. The last EMF reading in the first deter- mination is used as the first reading for the next determination. Thus three values of ac-dc difference are computed from 10 EMF readings. The three values are listed on the last line of data— 10, 0, 108, 1 17, 1 12, 4, 1 12—and these data are also printed on a TTY under appropriate columns. These figures

are, from left to right, the frequency in kHz, the ac-dc dif- ference correction (in ppm) to the standard TVC (computed

from an equation which was devised to simulate the correction curve for the standard in use), the three values of ac-dc dif- ference, the standard deviation of the three, and the average

of the three with the standard TVC correction applied. If the standard deviation is larger than 10 ppm, additional deter- minations are made. When one range is completed, parameters

for the next range are requested at the next line (Fig. 5) and

entered on the last line. If only one frequency is to be used, the

last entry on the line is 0.

Other Measurements

Measurements of n Measurements of n are made automatically by connecting

a TVC to the AC-DC 1 terminal and entering into the CRT the lowest and highest voltage to be applied. Tests are usually

made on a 100-V range at 10-V intervals from 60 to 100 V; however, this can be varied. Data obtained on one range of a multirange TVC are usable on any range if the values of n are

correctly related to output EMF of the TE. At each voltage

level the TVC output E is measured, and the potentiometer

is nearly balanced. However, a small imbalance is deliberately

programmed into the routine to obtain a negative DLA reading. The input voltage is then increased by a 0. 1 -V step so that

a positive reading is obtained and then reduced by the same

step for a repeat of the negative reading. This procedure is repeated three more times and four values of AE are computed. The average of the four is used as AE, and AV/V is computed with A V = 0.1. A value of n is thus computed at each voltage level using (3) and printed on the TTY along with

the voltage, EM F, and the DLA readings. The values of n are

plotted, a straight line is fitted to the upper two-thirds of the curve, and the parameters of that line are stored in the computer. Values of n can then be approximated by the computer

with sufficient accuracy by evaluating the straight-line

equation of any value of E.

In an alternative procedure, a certain percentage change in TE heater current is introduced by connecting a resistor in series with a TVC and alternately opening and closing a relay

which short circuits the resistor. A 40-0 resistor is shorted to introduce a 0. 1 -percent change at any voltage level in a 40-kQ

200- V TVC. This procedure can be used after merging thirteen substitute lines into the software described above.

Comparison and AC and DC Voltages

AC and DC voltages may be compared by connecting the ac voltage to the AC terminals (Fig. 4), with switch S] in the center or OFF position and an appropriate TVC at AC-DCl

AC and the two polarities of dc voltage are connected to the TVC by relays A: 1 and k 2

. The EMF £ of the TVC is measured,

and the potentiometer is balanced. Changes in the imbalance

EMF in the potentiometer circuit are computed from DLA

readings, and the difference between the ac and dc voltage is computed using equation ( 1 ), but with the TVC correction 5 applied. A dc calibrator may be connected to the DC terminals

(Fig. 4) and used as both supply and standard, or a separate standard (volt box and potentiometer or digital voltmeter) can be used at the DC-Std terminals. The standard should not be

affected significantly by changing the polarity of the applied

voltage. At very low voltages, where the voltage drop across the line resistance may be significant, it is necessary to monitor the dc

voltage at the DC-Std terminals. The cables with conductors

c and shields s (Fig. 4) are identical to equalize the voltage

634
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL IM-29. NO. 4, DECEMBER 1980

drop between these terminals and the comparison points at relay ki- These cables can be seen on the left rear table on the photograph of the console (Fig. 1 ).

Conclusion

A thermoelement comparator has been developed with a self balancing potentiometer and a divider, and it has been inte- grated into an automatic calibration system for measurements

of the ac+dc difference of a TVC relative to a similar standard instrument. The potentiometer is also used in automatic measurements of the n characteristic of thermoelements and

ac-dc voltage comparisons. The system requires an operator

to make the electrical connections and to select the ranges, as few if any TVCs have provisions for automatic range selection. The operator also enters the test parameters at a computer

terminal (see last two lines. Fig. 5). The system proceeds automatically beyond that point; however, it will stop at a number

of points if instability or other conditions prevent a DLA

reading or a balancing operation with a specified time. More expert maintenance is required than for manual sys- tems, but data transcription errors are very unlikely and data

storage for further reduction, pooling, analysis, and automatic

test report generation is feasible. As presently written, the software accommodates input instructions for only one or two

test frequencies; however, provisions could be made for a table of frequencies if a comprehensive scan were required.

The system saves time in the preliminary balances, fre- quency changes, and in data processing. However, due to the

settling time required for the thermoelements in the TVCs,

it proceeds at about the same rate as manual operations when

ac and dc voltages are being alternately applied to the instru- ments. A major advantage is that the operator is largely re- lieved of a task that is probably more tedious and fatiguing than those of most other types of calibrations.

References

[

1 ] F. L. Hermach and E S. Williams, "Thermal converters for audio fre- quency voltage measurements of high accuracy," IEEE Trans Instrum. Meas., vol. I M- 15, pp. 260-268, Dec. 1966

[2] E. S. Williams, "Thermal voltage converters and comparators for very accurate Ac voltage measurements," J Rijs. Nat. Bur Stand., vol. 75C,

pp. 145-154, 1971.

[3] C. H. Popenoe and M. S. Campbell, "Modular interactive data acquisition system—description and specifications," NBS Tech. Note 790, 1973.

[4] IEEE Std. Digital Interface for Programmable Instrum., IEEE Sid. 488,

1978.

635
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL IM-29. NO 4, DECEMBER 1980 An International Comparison of Thermal

Converters as AC-DC Transfer Standards

O. P. GALAKHOVA, S. HARKNESS, FRANCIS L. HERMACH, FELLOW, IEEE, H HIRAYAMA,

P. MARTIN, T. H. ROZDESTVENSKAYA, AND EARL S. WILLIAMS

Abstract—The ac-dc differences of 4 sets of thermal current and

voltage converters were determined at 40 Hz, 20 kHz, and 50 kHz with reference to the standards of the national metrology laboratories of

the United Kingdom, Japan, the Soviet Union, and the United States. For each voltage and current range and at each frequency, the average ac-dc difference determined by each laboratory for the 4 sets differed from the average of all laboratories by less than 10 ppm.

Introduction

THERMAL converters (each consisting of a conductor

heated by a current and one or more thermocouples to sense the temperature rise) are the basic standards for all ac

current and voltage measurements at audio frequencies. Extensive research in several national laboratories has indicated

that ac quantities can be determined in terms of dc standards

to 10 ppm or better with converters of proper design and construction [1]. Such accuracies are needed to support recent developments in electronic instruments and calibrators. New

types of multijunction thermal converters (MJTC's) show

evidence of even better accuracies.

International Tests

Because of the importance of converters as fundamental

standards of ac-dc difference, an international comparison of

the standards of four national metrology laboratories (Institute of Metrology (IMM), Leningrad, USSR; Electrotechnical Laboratory (ETL), Tokyo, Japan; National Bureau of Stan- dards (NBS), Washington, DC and National Physical Laboratory (NPL), Teddington, UK) was agreed upon at the 1968 meeting of the Consultative Committee on Electricity (CCE)

of the International Committee on Weights and Measures. NBS served as the coordinating laboratory. Each laboratory furnished a set of five converters with ratings (ranges) of 10 and 30 mA and 10, 30, and 100 V, and evaluated all 4 sets at 40 Hz, 20 kHz, and 50 kHz. (Regrettably, NBS inadvertently omitted making 40-Hz measurements.) The ac-dc differences

6 were reported for each converter. [5 = (Qac — (?dc)/(?dc

Manuscript received July 21, 1980. O. P. Galakhova and T. H. Rozdestvenskaya are with the Mendeleev All Union Institute of Metrology (IMM), USSR

S. Harkness and P. Martin are with the National Physical Laboratory (NPL), UK.

F. L Hermach and E. S Williams are with the National Bureau of Stan- dards (NBS), Washington, DC.

H. Hirayama is with the Electrotechnical Laboratory (ETL), Japan

TABLE I Estimated Overall Uncertainties h in ppm (Including the

Standard Deviation of the mean c and the Systematic Error <)

Laboratory Quantity

u (ppm) 40 Hz - 20 kHz 50 kHzSee Note f

IMM voltage 15 15 1

ETL current 10 10 2 voltage 15 25 2

NBS current 5 7 3 voltage 10 15 3

NPL current 3 tot A it voltage 3 to 7 4 to 8 4

Notes:

(1) u - e + o (2) o < 1 ppm

(3) u - c + 3o. 3 o - 2 ppm

(4) o • 1 to 2 ppm

where QdC is the average of the two directions of the dc quantity

(current or voltage) required to produce the same output

Electromotive Force (EM F) as the ac quantity Q3C ] The tests were made by IMM, ETL, NBS, NPL, and IMM in that order. After the tests of each laboratory were completed, the

results of all previous tests were forwarded to that laboratory.Because of the time required in each laboratory for the construction and preliminary testing of the converters comparisons did not actually begin until 1 972. Because of the delays and damages in shipment and the pressure of other tasks, they were not completed until 1977.

National Standards and Measurements

General

Brief descriptions of the ac-dc transfer standards and of the methods and equipment for comparing them in each of the 4

laboratories are given in this section. The accuracies claimed

by each laboratory in the evaluation of the interlaboratory

(traveling) converters are shown in Table I.

National Bureau of Standards

The NBS primary standards for ac-dc transfer at frequencies up to 50 kHz consist of a group of single-junction ther- moelements (TE7 s), with Evanohm or Karma heaters to

636
GALAKHOVA el al.\ THERMAL CONVERTERS AS AC-DC TRANSFER STANDARDS

minimize Thomson errors (which are calculated to be less than

2 ppm), and MJTC's [2]. Each thermal voltage converter (TVC) consists of one or more thin-film cylindrical resistor, mounted coaxially in a metal cylinder and connected in series with a 5-mA TE. Twelve

resistors give ranges from 1 to 1000 V [2]. With 2 TE's, ad- jacent ranges may be compared in a step-up or build-up process. The resultsliave verified that reactance errors (which can

be calculated roughly from the geometry) are less than 2 ppm

for ranges below 200 V. The ac-dc differences of higher ranges

are assigned from these comparisons.

To increase the precision of ac-dc current comparisons, the

heaters of two TE's are connected in series, and the outputs are connected to an "EMF comparator" which balances the 2 outputs [2]. Changes in input current cause proportional

changes in the EMF's, with little resulting change in the bal- ance. The ac, dc, dc reversed, and ac currents are applied in succession for equal time intervals to reduce errors from drifts

in the TE's, and each current is adjusted to obtain the same

output from one TE. The difference between the 2 ac-dc dif- ferences b\ — 5j is determined from the resulting readings of

a balance detector (nanovoltmeter). To compare TVC's in the same way, their input circuits are connected in parallel. The

NBS interlaboratory standards are similar in construction to the basic ac-dc standards.

National Physical Laboratory

The NPL comparisons are made using a bridge circuit similar to that used at NBS. A semiautomatic system was

developed, the switching between ac, dc forward, and dc re- verse being made by mercury-wetted relays in an automatically

timed sequence. A single measurement of ac-dc difference is obtained from 10 measurements of the bridge output when

successively energized from ac, dc forward, dc reverse, etc. The

bridge output is measured by an N PL-modified photocellgalvanometer amplifier to form a high input-resistance volt- meter having a most sensitive range of ±1 fiV, a linearity of ±0.2 percent, and a peak-to-peak noise of 4 nV with a 250-

source. The output is recorded using conventional data-logging

equipment. This data is analyzed by computer, using a curve-fitting technique [3], to determine an ac-dc difference

relatively unaffected by nonlinear drifts present during the measuring period. Over the range 0.5 to 100 V the NPL maintains its ac volt- age capability by means of build-up measurements using re- sistors of 200, 900, 3000, and 10 000 Q in conjunction with

separate 5- and 10-mA thermal converters, each adjusted to have 100-Q heater resistance. The linearity of all resistors, which is vital to the buildup, has been checked using third- harmonic testing. On theoretical grounds it is assumed that the 900- resistor has negligible frequency error. The thermal converters have platinum leadouts, rather than

the conventional Dumet, to avoid ac losses which would cause

errors in high-frequency low-voltage measurements. With the exception of the 30-mA converter, which has a Nichrome

heater, all other converters have Evanohm heaters. The latter were constructed by a method which is believed not to impair

the Thomson coefficient of the heater material. These converters were compared with the NPL Wilkins-type MJTC's,

and those with the best relative transfer errors were selected. The MJTC's were assigned zero transfer error at low and

medium frequencies.

The NPL interlaboratory standards consisted of 10-mA and

30-mA thermal converters for current measurements and a 5-mA thermal converter to be used in conjunction with 2000,

6000, and 20 000- Q resistors to provide voltage ranges of 10, 30, and 100 V. The higher value resistors were known to have

a voltage-dependent frequency response, believed to be due to the dielectric loss of the protective coating. A frequency-independent error of about 1 0 ppm, probably attributable to the Nichrome heater, was known to exist in the 30-mA converter.

Electrotechnical Laboratory As reported previously [4], the ETL primary standards have been built up from 10-mA converters according to the fol- lowing procedures:

1) After preliminary experiments on single and multi- junction-type converters, the most appropriate construction was decided upon.

2) Three 10-mA single-junction vacuum-thermocouple

current converters (Nikrothal heater and Chromel-Constantan couple) were selected, considering their characteristics and stability with time. Applying a theoretical formula, the ac-dc differences for individual converters and

then their mean —5.1 ppm was calculated. This mean has been assumed to be constant up to 1 00 kHz. The ac-dc dif- ferences of other current converters were determined by

comparison with these 3 standards. This method is different from that for determining ac-dc difference in other national

laboratories. The effect of the assignment of —5.1 ppm is partially evident in the results of the international comparisons.

3) The ac-dc differences of 30-mA converters were deter- mined by comparison with three 10-mA converters in parallel. 4) The frequency dependencies of 1-kfl, 3-kfi, and 100-kf2

series resistors were determined on the basis of a calculable

single-wire resistor. Then the ac-dc differences of 10, 30, and 100-V voltage converters and their frequency dependencies were determined by combining the ac-dc difference of the 10-mA current converter with the frequency dependencies of the resistors. For ac-dc measurements, a comparator [4] of 0.1 -ppm

resolution and accuracy better than 1 ppm up to 100 kHz is used. The standard deviation of a single measurement is nor- mally better than 1 ppm.

For the international comparison, detailed measurements

at 40 and 1 592 Hz and at 20, 50, and 90 kHz were made on the ETL interlaboratory standards during 1973, and measure- ments to check their stability with time were made in 1978. The stability was within 10 ppm for 5 years except for converters with faults. From ETL's experience, the development

of converters more resistant to shocks and vibration during

transportation is desirable.

637
II I I TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT. VOL. IM-29. NO. 4. DECEMBER 1980

Mendeleev All Union Institute of Metrology

For the comparisons of current converters at IMM, a standard which reproduces the unit of ac current in the fre- quency range from 40 Hz to 100 kHz is used [5]. For the comparisons of voltage converters, a standard which repro- duces the unit of ac voltage in the frequency range from 20 Hz

to 30 MHz is used [6]. The main elements of these State Standards include specially developed and studied noncontact

current and VTC's [7]. The voltage standard comprises 8 low-frequency (20 Hz to 100 kHz) and high-frequency ( 100 kHz to 30 Mil/) converters with nominal values of voltage

from 0.3 to 10 V. The units of ac current and voltage arc realized at different frequencies, using the Slate Standards, by the method of se- quential comparison of ac current (or voltage) with dc current

(or voltage) which produces an equivalent effect in an appropriate thermal converter. Theoretical analysis of the method

and investigation of these standard thermal converters, de- veloped at I M M, have been carried out. They showed that the equality of the thermal EM Fs, generated by a converter, when

its heater is alternately connected in dc and ac circuits, cor- responds with high accuracy to the equality of the effective value of the measured ac current (or voltage) and dc current

(or voltage) [8]. Each of the N BS, N PL, and ETL interlaboratory current and voltage converters was compared with an IMM standard

converter by connecting the two converters in the ac circuit, then in the dc circuit whose polarity was then reversed, and

then in the ac circuit again. Each time the thermal EMFof an

interlaboratory converter was measured by means of a dc potentiometer, the thermal EMFof the IMM standard converter being kept constant. The error of the transfer was determined

as the difference of mean values of the thermal EM F's of the interlaboratory converter obtained when the converter was

connected in the ac and dc circuits.

Results and Disc ussion

The ac-dc differences in ppm assigned to each interlabo- ratory converter by each laboratory in terms of its national standards are given in Table II. The most significant result of an interlaboratory comparison

is an indication of the agreement among the laboratories. In the present comparison, this is most readily shown by averaging, for each laboratory, at each frequency, the results of the

tests of all 4 converters of a given range, since the effects of random errors and changes in individual converters are thereby

reduced. These averages arc shown in Fig. I . Because the ETL

converters were not available then, none of the results of the

first lest at IMM arc included in the figure or in Table II. The converters were reasonably stable during the tests in any one laboratory. However, shipment did cause problems:

ETL had to return the NPL TVC's for repairs, and IMM

noted a 10-ppm change in its 100-V converter at 50 kHz be- tween its first and second tests. The change probably occurred

in transit after the NPI. tests. There may also have been a change in the 10-mA converter after the first IMM test. The

average change for all of the other converters between the two IMM tests was less than I ppm. fi"}o

TABLE II The: AC-DC Differences in ppm of each Converter as Determined by Each Laboratory

MEASURINC LABORATORY INTERLAI

ETL NHS HPL IMM STANDARD RANCE FREQUENCY

0. / 0.9 8 HPL 10 V 40 Hi

- 5.1 - 5.9 3 en.

- 8.8 -12.0 0 IMM

- 3.4 - - 4.7 4 ins

0.0 0.3 9 NFL 30 V 40 Hi

- 4.3 - 5.6 6 ETL

- 8.6 -13.0 0 IMM

- 4.0 - - 4.1 N7JS

4.3 - - 0.5 8 HPL 100 V 40 Hi

- 4.8 - 8.9 3 ETL

- 5.1 - 9.5 0 IMM

- 0.1 - - 4.9 5 NBS

- 5.8 - - 5.4 NPL 10 uA 40 Hi

- 4.9 - - 4.5 5 ETL

0 IMM

- 2.5 - 2.8 15

-14.6 _ -11.1 -9 HPL 30 mA 40 Hi

- 7.8 - - 4.6 -3 ETL

- 6.0 - 1.8 0 IMM

- 7.2 - 2.3 -1 NBS

- 4.5 (i - 0.8 7 NPL 10 V 20 kHz

- 4.2 - 2 - 3.7 6 ETL

- 4.5 - 4 - 5.9 0 IMM

- 1.7 - 1 - 1.1 6 NBS

- 6.3 - 2 - 1.9 7 NPI. 30 V 20 kHi

- 5.5 0 - 3.9 7 ETL

-10. 2 - 7 - 9.7 0 IMM

- 4.6 - 1 - 1.4 7 NRS

- 4.0 - 7 - 6.0 3 NPL 100 V 20 kHi

- 4.8 - 4 - 7.0 6 ETL

- 6.4 - 9 - 9.3 0 IMM

0.1 - 2 - 1.4 7 NBS

- 2.5 - 2 - 3.4 -3 NPL 10 mA 20 kHi

- 5.1 - 1 - 4.3 -3 ETL

- 1.9 4 2.1 0 IMM

- 2.8 - 2 - 2.5 4 NBS -15.6 -1J -11.1 -9 NPL 30 mA 20 kHi

- 8.8 - 4 - 4.5 -5 ETL

- 5.6 0 1.8 0 IMM

- 7.2 0 - 1.5 /

- 5.3 2 0.0 -5 NPL 10 V 50 kHi

- 3.7 - 2 - 1.9 -4 ETL

6.1 6 4.1 0 IMM

- 0.4 3 - 0.1 -4 NBS

-10.5 - 3

- 3.7 1 NPL 30 V 50 kHi

- 6.0 1 - 1.5 i. ETL

- 9.6 - 4 - 7.1 0 IMM

- 4.9 0 - 0.6 5 NBS

-15.2 11 -14.2 -10 NPL 100 V 50 kHi

- 6.2 - 2 - 7.3 1 t n. -11.0 -12 -11.4 -10 IMM

- 0.3 - 2 - 1.3 4 NBS

- 4.6 - 3 - 1.6 -4 NPI 10 mA 50 kHZ

- 5.6 0 - 3.3 -5 ETL

- 3.5 8 8.8 0 IMM

- 4.0 0 - 1.9 4 NBS

-16.0 14 -11.0 -1" NPL 30 mA 50 kHi

- 9.0 1 - 5.5 -5 ETL

- 6.9 (1 1 1 0 IMM

- 8.0 0 - 1.0 -2 NBS

As shown in Fig. 1, the differences between the averages

obtained by the laboratories for the 4 converters of a given range at a given frequency were very small. The largest dif- ference between any one of these laboratory averages and the mean of the 4 laboratory averages at the same range and fre- quency was less than 9 ppm. The average of these differences (without regard to sign), taken over all laboratories, ranges, and frequencies, was less than 3 ppm.

Since the same 4 converters of each range were evaluated

at each laboratory, it is possible to test the hypothesis that any

offset (systematic difference) between 2 laboratories is masked

by the imprecision of the measurements; i.e., that the differ-
GALAKHOVA el ai: THERMAL CONVERTERS AS AC-DC TRANSFER STANDARDS

- 0 J -10

L»t»f»lOfl»0 40 Hz 20k Hi 50 k Hz S ftl till till I

•100V

30mA

Fig. I . Average ac-dc differences in ppm as determined by each of the 4

laboratories.

ences between the averages shown in Fig. I are not significant because of the random differences between the results of tests of individual converters. For 4 laboratories there are 6 such

calculations for each range and frequency. To reduce the number, one laboratory can be used as a reference. Experience indicates that the random error in comparing

converters is not strongly dependent on range or frequency but may differ for different laboratories. There are also random

changes in the converters between tests by two laboratories. On the basis that they are also roughly independent of range and frequency, the precision of testing the hypothesis can be

increased by computing a pooled standard deviation s0 for each

pair of laboratories from all of the comparisons between them.

If Si is the standard deviation of the differences between the

pairs of measurements in two laboratories for the 4 converters

of a given range at a given frequency, and there are k such

subsets of n = 4 pairs each, then

sl =

kEst

TABLE III Evaluation of Differences Between Laboratories with NPL as Reference

Evidence of Offset

40 Hz 20 kHz 50 kHz RanRe ETL IMM ETL IMM NBS ETL IMM NBS

10 V yes yes

30 V yes yes yes yes

100 V yes yes yes yes

10 OA yes yes

30 mA yes - yes yes

The results of a series of such calculations are shown in Table III, with NPL as the reference. NPL was chosen because

its results are generally centrally located in Fig. 1 and because

it made measurements at all frequencies. A "yes" in Table 1 1

1

indicates that Iso/yfn < \da \, where d0

is the difference be- tween the averages obtained by the 2 laboratories for the

subset, because we can then decide that there is an offset

(statistically significant difference), and we reject the hypothesis that it is masked by measurement imprecision. A dash

indicates that iso/y/n > \ da | , so that there is no evidence of an offset. Although the table shows offsets in about one third of the cases, it should be emphasized that the differences be- tween laboratories actually observed were much smaller than

the assigned uncertainties shown in Table I. 63!

The pooled standard deviations sq were 2.10 ppm for ETL-NPL, 2.59 ppm for IMM-NPL (k = 1 5), and 2.07 ppm

for NBS-NPL (k = 10). The distributions of the differences between NPL and the other laboratories were reasonably normal. In only one case did the pair of tests on an individual converter in 2 laboratories differ from da by more than 3^o. possibly indicating a change in that converter.

Conclusions

It is apparent that the agreement among the laboratories

is much better than the uncertainties estimated in Table I. In most cases the differences between the averages obtained by

the laboratories are not statistically significant because of the

scatter of the measurements. The primary ac-dc transfer standards of the 4 laboratories are all thermal converters, but

are of different types of construction. The methods and

equipment for comparing converters are also different. Further, the primary converters were studied independently and

thoroughly in each laboratory. It is extremely unlikely that all of them could be afflicted with an ac-dc difference of the same

magnitude and sign. The excellent agreement in these careful

international tests provides valuable assurance of the absence

of significant systematic errors in the primary standards and

the comparison apparatus. It is thus highly significant. There

is now a very firm basis for international concordance of the ac-dc transfer to better than 10 ppm at audio frequencies.

References

[1] F. L. Hermach, "AC-DC comparators for audio frequency current and

voltage measurements of high accuracy," IEEE Trans. Insirum. Meas.,

vol. IM-25, pp. 489-492, Dec. 1976.

[2] F. L. Hermach and E. S. Williams, "Thermal converters for audio-fre- quency voltage measurements of high accuracy," IEEE Trans. Insirum. Meas., vol. 1M-15, pp. 260-268, Dec. 1966.

[3] S. Harkness and R. B. D. Knight, "Computer correction of errors due

to instrumental drift by curve fitting," Natl. Phys. Lab., Eng. DES Memo

No. 20, Nov. 1975.

[4] S. Iwamoto and H. Hirayama, "AC-DC thermal converters of the ETL," IEEE Trans. Insirum Meas., vol IM-23, pp. 326-329, 1974.

[5] O. P. Galakhova, E. K. Dranishnikova, and M. S. Belayeva, "Special state standard of the AC current unit in the range from 40 Hz to 100 kHz,"

Izmer. Tekh., pp. 42-44, Mar. 1976.

[6] T. B. Rozhdestvenskaya R. F. Aknaev, and O. P. Galakhova, "Special

state standard of the AC voltage unit in the range of frequencies from

20 Hz to 30 MHz," Izmer. Tekh., pp. 44-46, Mar. 1976.

[7] R. F. Aknaev, O. P. Galakhova, and T. B. Rozhdestvenskaya, "Methods

and instruments of assurance of the uniformity of AC voltage measurements," Izmer. Tekh., pp. 66-71, Apr. 1976.

[8] R. F. Aknaev and T. B. Rozhdestvenskaya, "Analysis of errors introduced by thermal converters when comparing AC and DC current and voltages," Trudy Metrology Inst. SSSR, 138(198), 1972.
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-29, NO. 4, DECEMBER 1980 A Semiautomatic System for AC/DC Difference

Calibration

KEITH J. LENTNER, member, ieee, CLIFTON B. CHILDERS, and SUSAN G. TREMA1NE

Abstract—A semiautomatic ac/de difference calibration system is described. The system operates over a frequency range of 20 Hz to 100 kHz, covering the voltage range from 0.5 V to 1 kV. For all voltages

at frequencies in the range from 20 Hz to 20 kHz, the total uncertainty

is 50 ppm and 100 ppm for voltages at frequencies between 20 kHz and 100 kHz. In addition to ac/dc difference testing, the system can be

readily adapted to calibrate precision ac digital voltmeters or ac calibrators. Results of extensive intercomparison testing of the new system

against a manual test system, using a multirange thermal transfer instrument as a transport standard, are reported, and the results in- dicate that the differences obtained are well within the combined total uncertainty limits of the two systems.

Introduction

TECHNIQUES for performing precision ac/dc difference

testing, not only at the National Bureau of Standards

(NBS) but also in other national and industrial or govern- mental metrology laboratories, generally make use of manual

testing methods which utilize photo-cell preamplifiers and

light-beam galvanometers as voltage detectors, in conjunction with manually balanced voltage comparators [I], [2]. Careful

attention to the elimination or reduction of systematic and

random uncertainties in testing methods and thermoelements

(TE's) has resulted in sufficient confidence in test data to allow

results to be reported with total uncertainties at the 10-100

ppm level (or better, in some cases) over wide voltage and

frequency ranges [3], [4]. These manual test methods, however, are very time consuming and subject to errors due to operator fatigue. To overcome these deficiencies, and to ex- pedite test report generation, a semiautomatic ac/dc cali- bration system was developed. Because of the low-level voltages that must be resolved (10

nV or less), large temperature coefficients of TE's, ac or dc

source instabilities, and dc reversal differences of TE's,

stringent requirements are placed upon the measurement

system. In addition, light-beam galvanometers suffer the disadvantage of not being easily adaptable to automation.

However, with the advent of digital electronics, low-level

digital voltmeters (DVM's) are available with resolution and

accuracy sufficient to meet the requirements of precision au- tomatic ac/dc difference testing. Programmable ac and dc

voltage calibrators are also available which lend themselves

to adaption for automatic testing. Advances in analog/digital

Manuscript received July 24, 1 980. This work was partially funded by the U.S. Department of Defense Calibration Coordination Group. The authors are with the Electrosystems Division, National Bureau of Standards, Washington, DC 20234.

interfaces have greatly improved the ease with which computers can be used to control these instruments. The system, to be described in this paper, was designed to overcome the deficiencies of manual test methods by taking

advantage of the electronics advances discussed above.

Specifically, some objectives were to: 1) use commercially

available programmable instruments whenever possible; 2) eliminate the need of manual data entry into a computer for data reduction; 3) eliminate the frequent data checks and cross checks required in manual tests; 4) eliminate as much operator

intervention as possible during the testing procedure; 5) closely control the time interval between application of both ac and

dc voltages to the test circuit in order to minimize drift effects; 6) provide as much computer-operator interaction as possible

to simplify the testing procedure; and 7) achieve the same level of uncertainty as is presently possible with manual testing methods. The system operates over a frequency range of 20 Hz to 100 kHz, covering the voltage range from 0.5 V to 1 kV. For all voltages at frequencies in the range from 20 Hz to 20 kHz, the

total uncertainty is 50 ppm and 100 ppm for voltages at frequencies between 20 kHz and 1 00 kHz. In addition to ac/dc

difference testing, the system can be readily adapted to cali- brating precision ac digital voltmeters or ac calibrators.

System Hardware

The hardware implementation emulates that used in manual

systems at NBS. The prinicpal instrumentation difference is the use of an electronic voltage detector. Also, almost all the instruments are programmable. The major problem in hard- ware development was obtaining low-thermal-noise switches which could be automatically controlled. TE voltages of about

10 mV need to be measured to within 1 ppm (10 nV), and

solid-state relays were found to be unsuitable since most of these devices have thermal noise of the order of 1 ^V. The

problem was solved by modifying manually operated low- thermal-noise switches. Solenoids with 24-V operating coils were mechanically linked to the low-thermal-noise switch, and

are energized by programmable relays. Experience gained at NBS throughout many years of manual testing was used to design the remainder of the system. The switches which apply the ac or dc voltages to the test cir- cuit are relays capable of switching up to 5 kV. The 24-V op- erating coils of these relays are supplied in turn by means of programmable relays.

640
LENTNER el at: SYSTEM FOR AC/ DC DIFFERENCE CALIBRATION

CRT and

Keyboard

(Syelem

Console)

16-Bit Minicomputer System

BS 232C I lnl»ftlc» Modular Interactive Oata Acquisition System (MIDAS)

Programmable DC Standard

TTY

(Test Report)

NBS

Range

Resistor NBS

Thermoelement Standard

>C ... DC

»C ' Y » A

..Programmable iSOl £*""" AC Standard

(5=

j > AC/DC > | Transfer i NBS TE f

Standard: Comparator f~gh or Unit

| (K-V Divider & 7~^f Under

Potentiometer) I Test Low Thermal

Switches

Digital Voltmeter and

Null Detector

BCD Intarlac*

Fig. I . Simplified system block diagram.

A simplified block diagram of the system configuration is shown in Fig. 1 , and a photograph of the system in Fig. 2. The

16-bit minicomputer serves as the system controller and has 64 kbytes of core memory with a dual floppy-disk system for program and data storage. The system console (CRT), the

teletypewriter (TTY), and the Modular Interactive Data

Acquisition System (MIDAS) [5] are connected to the computer by means of interfaces which conform to the EIA RS232-C [6] standard. MIDAS is a user-oriented digital interface system for data acquisition and experiment control. Its hard- ware is based on the CAMAC [7] standard and is composed

of several functional modules which may be plugged into slots

in a standard crate. A MIDAS power supply and controller are the only two additional items of equipment necessary to make MIDAS operational. Communication between the MIDAS controller and individual modules is in accordance

with the ASCII [8] standard code for information inter- change. As indicated in Fig. 1 , the programmable ac and dc sources and the voltage detector are controlled by binary-coded-decimal (BCD) MIDAS modules. The sequence of applying

voltages to the test circuit, as indicated in Fig. 1, is ac, dc+,

dc— , ac, and the ac or dc source voltage relays are controlled bv means of a MIDAS relay module. The choice of this par- 641
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 1M-29, NO. 4, DECEMBER 1980

ticular voltage application sequence has been fully discussed

in reference [2] and will not be reiterated here. The NBSdeveloped TE comparator (consisting of a Kelvin-Varley

voltage divider and Lindeck potentiometer) has been described

in the same reference. The low-thermal-noise switches, controlled by a MIDAS relay module, are solenoid actuated

switches which select the proper voltage output from the TE

comparator. The detector obtains readings of the proper

voltages and passes these readings in BCD format to the computer where they are stored for later processing. The

voltage detector is a digital voltmeter which has a steady-state

input impedance of about 10 Mfi and is capable of 1-nV res- olution on its most sensitive ( IO-jiV) range. The Kelvin-Varley

divider in the TE comparator is used to adjust the difference between the output voltages of the standard TE and the unit under test (UUT) to within 500 nV. A typical programming sequence will be described in more

detail in the next section. For simplicity in Fig. 1 , two digital

panel meters which monitor the voltage outputs of the dc and

ac programmable standards are not shown. Also not shown is a counter used to monitor the ac standard's frequency. This

equipment is visible in the system photograph (Fig. 2). The

large instrument on the left of the test console table is an ac

calibrator under test. As was mentioned previously, the system can be readily

adapted as well to the tasks of calibrating precision digital voltmeters or ac calibrators. In this case, the testing procedure

and balance equation differ slightly from those described in reference [2]. For example, the corrections to the nominal

output voltage dial settings of an ac calibrator can be deter- mined by using the system's dc programmable standard volt- age source (set to the same nominal value as the ac calibrator)

and a thermal voltage converter (TVC), both having known

corrections. For this case, only the Lindeck potentiometer

portion of the TE comparator is used. The correction to the UUT in ppm (whether that of an external ac calibrator or the system's programmable ac standard) is given by

ns Vs where b5 is the ac/dc difference correction of the TVC at the

test frequency, Va

is the reading of the detector (DVM) with

ac voltage applied (i.e., the difference between the TE's output

emf and the fixed voltage of the Lindeck potentiometer), Vj

is the average of the detector readings with direct voltage of both polarities applied, ns is a dimensionless factor equal to 2 if the TE has a square-law response, Vs is the standard TE's

output EMF at the applied test voltage, and Cdc is the cor- rection to the dc voltage standard's nominal output.

System Software

In order to reduce the effort involved in making routine tests of TE's and TVC's, the software for the system was designed

to provide as much interaction between the system console (the CRT) and the operator as possible. This interaction was used

to reduce the possibility of damage to either the standard TVC

or the UUT when either software or hardware failures occur.

Also, by progressing logically through the first portions of the

test routines, the operator, when necessary, can take whatever

corrective actions might be required. A significant advantage

of the software design is that sufficient warm-up time is au- tomatically provided between application of the various test voltages so that the effects of errors due to the warm-up drifts of the TE's are reduced. In addition, the computer can closely control the time intervals between DVM readings; hence, the

effects of linear drift in the ac or dc sources are minimized.

The programs written in Basic language for this system

provide the means for automatically setting the ac or dc source voltages and ac frequency, operating the voltage relays and

low-thermal switches, triggering the DVM to obtain voltage

readings, and storing these readings in the computer memory. About thirty programs have been written with six constituting the major testing software package. Program ACDCLV is used for making ac/dc difference tests at voltages from 0.5 to 95 V, and ACDCHV is used for similar tests at voltages

greater than 1 05 V up to 1 kV. NTSTF is used to determine

a TE's dimensionless factor n [1], [2] and includes provision

to perform a linear least squares fit of the experimental data. ACCAL is a program used to calibrate an ac calibrator, a

precision ac-digital voltmeter, or the ac source of the system

in terms of a standard TVC and the system's calibrated dc

supply, as discussed in the previous section. Stability of the system's dc and ac sources can be checked

by using programs called STABLV and STABHV. These programs are used at voltage ranges similar to those for ACDCLV and ACDCHV. These stability tests are useful to determine whether or not the supplies are functioning properly.

Occasionally, test results appear to be erratic and the stability

tests are a convenient method to ascertain whether the erratic results may be due to the ac or dc sources. Stability of the sources is measured by using a programmable 6'/2 digit DVM

whose function, ranges, and reading rates are remotely controlled.

Initial stability test parameters for voltage, frequency, time between readings, and total number of readings are entered

into the computer. The program then exercises complete au- tomatic control over the ac and dc sources, the voltage relays, and the DVM. At completion, stability data including the values of minimum, maximum, range, average, standard deviation of the mean, and three times the standard deviation of

the mean for both ac and dc voltage readings are printed on

the TTY.

Operation

Since the major application of the work described in this paper has been directed toward ac/dc difference testing, a

typical sequence of operations for this type of testing will be

described. ACDCLV is a program which provides partial

automatic control of ac/dc difference testing. The program

is "partially automatic" in that an operator must manually

input test parameters, switch the TE comparator to select proper TE's, and must manually balance the TE comparator's

Lindeck potentiometer and Kelvin-Varley voltage divider. Basically, ACDCLV contains three general sections. The

first section provides program initialization and documenta642
LENTNER ci aU SYSTEM FOR AC/DC DIFFERENCE CALIBRATION

tion. The midsection includes a network of subroutines that

interact with the operator. Instructions displayed on the CRT

prompt the operator to: 1 ) key in test parameters which set ac and dc voltages and frequency; 2) switch ihe TE comparator

to determine whether the standard TVC or UUT has the

greater TE output, ensuring that the TE's are connected to the proper input terminals of the TE comparator; 3) balance one TE's output with the Lindeck potentiometer; then 4) balance

the Kelvin-Varley divider. Repeatedly accessed subroutines

then close and open the appropriate relays, and obtain the DVM voltage readings for four determinations of ac/dc dif- ference at each test point. Finally, in the third section, statistical calculations are performed, and the results are printed on the TTY. Options are provided, and the operator can choose whether to stop, repeat a test, or change parameters (either voltage or frequency) and run another test. This program is used for voltages <95 V. Since the MIDAS interface does not have isolated analog

and digital grounds, it was necessary to develop a separate software package for high-voltage testing. At high voltages,

it was found that the opening or closing of either the ac or dc

voltage relays introduced transient voltage surges into the MIDAS digital circuitry. The MIDAS controller interpreted

these surges as voltage level changes identical to programmed

input commands. The result was that incorrect relays might

be opened or closed, the voltage detector might be falsely triggered to obtain a reading, or any other of a series of un- wanted events might occur. Hence, the ACDCHV program

was developed to ensure that no high-voltage supplies are turned on whenever a voltage relay is opened or closed. The

specific procedures include;

1 ) Input test number, voltage, frequency, observer's name,

and code for the standard TVC used.

2) Visually check to see that both software and hardware

are operating correctly by checking the system's digital panel meters and frequency meter for correct readings of the outputs

from the ac and dc sources. If the outputs are incorrect, try again.

3) Manually set the TE comparator to its high TE voltage

position and switch the voltage detector to its 10-mV range. The program automatically closes the dc voltage relay and a low-thermal-noise switch, turns on the dc supply, and triggers the detector to obtain the TE voltage output reading.

4) Switch the TE comparator to its low TE voltage input

position.

The program then commands the detector to read the voltage.

The values of the high- (from the previous step) and lowvoltage readings are compared. If the values are incorrect, the operator is instructed to interchange the two TVC inputs after the dc voltage has been turned off. The program automatically

stores the value of E\ovl, since that value is used in the balance

equation [2, appendix].

5) Switch the TE comparator to the high TE input position, turn on the Lindeck potentiometer, change the detector to its 10-jiV range, and manually balance the potentiometer.

6) The first low-thermal-noise switch is opened and a sec- ond closed. The operator now manually balances the Kelvin- Varley divider for a near null reading on the detector. The dc

source is automatically turned off, and the dc voltage relay

opened.

7) The ac voltage relay is closed, the ac supply turned on, and its voltage is automatically incremented or decremented

until the UUT's TE output voltage is < 1 ^V of its value with dc voltage applied as in step 6. 8) At this point, the manual initialization process is complete, and the test proceeds automatically under program

control. A sequence of near null voltage readings is obtained by the null detector when ac, dc+, dc— , and ac voltages are applied to the standard TVC and UUT. These four voltage

readings are each obtained from the average of ten readings

of the DVM. The standard deviation of the ten readings is computed and, if this value is <200 nV, the program continues.

If the standard deviation is >200 nV, due to drifts in either the standard TVC or the UUT, or the ac or dc sources, or any of

these effects combined, the series of ten readings is repeated. After five attempts, if the DVM readings have not stabilized, the test is aborted.

9) The sequence of readings with ac, dc+, dc— , and ac

voltages applied constitutes one determination of the ac/dc

correction to the UUT. The process is continued for four de- terminations; the average value of four determinations is computed and printed on the TTY. The operator then may

elect to repeat the test, change voltage range or frequency, or stop the test. The system was intended to calibrate high-quality TVC's,

and it was not necessary to add additional balancing steps to eliminate uncertainties due to large dc reversal difference (i.e., difference between the dc direct and reverse voltages required

for the same TE EMF output of the UUT). For the worst case, using 2.0 and 1 .6 as the two extremes of the characteristic n which may occur for the standard and UUT TE's, it can be shown [4] that systematic uncertainties of less than one-fourth

the UUT's dc reversal difference will occur. Minor software changes can eliminate this systematic source if large reversal differences are anticipated. The changes would amount to adjusting the ac voltage (cf. step 7 above) for a TE output

equal to that when dc reversed voltage is applied to the test circuit. The only penalty for such a change in the operating

procedure would be an increase in testing time of perhaps one

or 2 min (for four determinations of ac/dc difference). There is also the possibility of a systematic uncertainty if the ac or dc voltage sources drift during a sequence of four determinations. However, for the same extremes of «, the uncertainty is only 10 ppm if either source drifts by as much

as 100 ppm. The STABLV and STABHV programs readily

provide information about the stabilities of the sources. Measurement Results and Uncertainties

There are many error sources which determine the uncertainty of ac/dc difference measurements of the type described

in this paper. Similar manual systems have been in operation

for many years at NBS and their uncertainties have been documented [l]-[3]. DC reversal difference of a TE, self- heating and ambient temperature effects, drift effects due to nonequal time intervals during the voltage measurements,

unbalanced lead impedances, and induced voltages from

643
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-29, NO. 4, DECEMBER 1980

TABLE I R ANCiF 01 AC/DC Differences for Transport Standard (ppm)

Test Test Frequency [Mi)

Voltage

Range VoltageVolts

20 100

0.25-0.5 0.5 18 19 23

0.5 - 1 1 9 2 It 1-2 2 1 6 9

2 - It 3 2 0 0

It - 8 e 0 l 3

8 - 16 16 8 15 3

16 - 32 30 8 10 9

32 - 6I4 SO 3 3 10

6U - 125 96 ? 6 12

125 - 250 200 It 2 lit

250 - 500 500 U 50 39

500 - 1000 500 2? 51. lit?

500 - 1000 1000 17 TO

electromagnetic fields are some of the more common sources

of error. The process of determining an estimated total uncertainty for a system is based upon experience with the sys- tem, careful measurements of standards with accurately known

corrections, or intercomparison tests of two independent test methods [3]. The latter procedure, involving comparison of

a well-characterized manual system with the new one, was

followed, since the uncertainty sources are believed to be the same in both systems.

The system performance was verified by an extensive testing

procedure which involved the use of a multirange thermal

voltage converter instrument as a transport standard. The

transport standard was first tested for ac/dc difference cor- rections at various combinations of applied voltages and

frequencies using the MIDAS-based system. It was then

tested, at similar voltages and frequencies with the same

standard TVC, using the manual testing techniques which have been used for many years at N BS. A third test was performed,

again using the MIDAS system. Finally, the MIDAS system was disassembled, physically relocated, reassembled, and

tested for proper operation. Then the transport standard was

tested a fourth time, again using the MIDAS system. It should be noted that not all voltage range-frequency combinations were tested for all four tests. However, in those cases where

fewer than four tests were performed, the results indicated

sufficiently small differences between at least two independent

test methods to establish that further testing was not warranted. The results of these independent intercomparison tests are summarized in Table I. In evaluating these results, it must

be remembered that, for calibrating multirange instruments

like the transport standard, the manual test system has

uncertainties of the same magnitude as the semiautomatic

system. Hence, combined total uncertainties of the order of 100 and 200 ppm are possible depending upon the test fre- quency.

In Table I, the column listing "Voltage Range" refers to the input voltage range of the transport standard, the "Test

Voltage" is the actual test voltage applied, and the different columns for frequency are the actual test frequencies used. The

numbers listed under the frequency columns are the maximum

differences (in ppm) of the calibrated ac/dc difference cor- rections for the transport standard as determined for at least two, and up to a maximum of four, of the test conditions de- scribed above. That is, the numbers reported are the range (magnitude of the maximum minus the minimum) of the ac/dc

difference corrections calculated for each test voltage and

frequency. The total elapsed time between the first and final tests of the transport standard was about eight weeks. From the test results at 20 kHz, the average value of the range of ac/dc differences for all voltages is only 8 ppm. At 50 kHz the average value of the range is 1 9 ppm, and at 100 kHz,

where skin effects and stray capacitance effects predominate,

the largest difference (at 500 V) was 147 ppm. However, this large range is within the expected combined total uncertainty

of 200 ppm for the two systems. Even with this one range

greater than 100 ppm, the average value of the range at 100 kHz is still only 23 ppm. No intercomparison results were

obtained at 1000 V, 1 00 kHz since the high-voltage amplifiers

in both systems could not supply sufficient power at that voltage-frequency combination.

Conclusions

The results of the work reported in this paper indicate that

it is possible to measure ac/dc differences semiautomatically

with uncertainties about the same as those obtained with manual testing techniques. The major advantage of the system appears to be the elimination of most of the tedium involved

in manual testing, in which deflections of light beam galva- nometers must be observed and recorded. For future work, it appears that improvements in system

operation can be obtained by development of self-balancing

TE comparators. Beyond that, it may be possible, by taking

advantage of improvements in voltage detector resolution,

desktop computers, and system components whose operation

is compatible with new digital/analog interfaces, to make

readily available the hardware and methodology for improved

ac/dc difference testing techniques.

Acknowledgment

The authors are indebted to E. S. Williams for his many

valuable contributions to the work reported in this paper. He

designed and built the voltage and low-thermal-noise switch modules and provided much of the initial software which

eventually became part of the total programming package

required for system operation. R. E. Kleimann, S. C. Bailey, and W. C. Caterisano each contributed portions to some of the computer programs.

References

[ I ] F. L. Hermach and E. S Williams, "Thermal converters for audio-fre- quency voltage measurement of high accuracy," IEEE Trans. Instrum.

Meas.. vol. IM-15, pp. 260-268, Dec. 1966.

[2] E. S. Williams, "Thermal voltage converters and comparator for very accurate ac voltage measurements," Nat. Bur. Stand. J. Res., vol. 75C,

644
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT. VOL IM-29, NO. 4, DECEMBER 1980

pp. 145-154, Dec. 1971.

[3] F. L. Hcrmach. "AC-DC comparators for audio-frequency current and

vollagc measurements of high accuracy," IEEE Trans. Insirum. Meas.,

vol. IM-25, pp. 489-494, Dec. 1976.

[4] R. S. Turgcl, "A comparator for thermal ac-dc transfer standards," ISA

Trans., vol. 6, no. 4, pp. 286-292, 1967.

[5] C. H. Popcnoc, M. S. Campcll. "MIDAS Modular Interactive Data

Acquisition System —description and specifications," Nat. Bur. Stand.

Tech. Note 790. U.S. Gov. Printing Office, Washington, DC 20402 (Aug.

1973).

|6] "Interface between data terminal equipment and data communication

equipment employing serial binary data interchange," ElA RS-232-C,

Electron. Ind. Ass.. 2001 Eye St., N.W., Washington, DC 20006 (Aug.

1969).

[7] "IEEE standard modular instrumentation and digital interface system (CAMAC)—computer automated measurement and control," IEEE

Stand. 583—1975. IEEE, 345 E. 47 St., New York, NY 10017.

[8] "American National Standard Code for Information Interchange," ANSI

X3.4—I977, Amer. Nat. Stand. Inst., Inc., 1430 Broadway, New York, NY 10018.

645
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-27, NO. 3, SEPTEMBER 1978 An RMS Digital Voltmeter/Calibrator for

Very-Low Frequencies

HOWARD K. SCHOENWETTER, senior member, ieee

Abstract— A portable rms digital voltmeter (DVM) has been

developed at NBS to support vibration measurements over the ranges

of 0.1 to 50 Hz and 2 mV to 10 V. A self-contained calibrator provides

for self-calibration and may be used for calibrating other VLF

voltmeters. The calibrator basically consists of a Kelvin-Varley

divider fed by a reference voltage (either dc or sinewave generated by

a ROM-DAC combination). A multijunction thermal converter (MJTC) was selected as the sensing device in the rms/dc converter of

the DVM since its low ac/dc difference facilitates calibration of the ac calibrator. Factors affecting accuracy and response time are

analyzed. The DVM response time is 40 s for the lowest input frequency. Its accuracy (percent of reading) is 0.1 percent above 0.5 Hz and 5 mV and 0.2 percent below these values. The ac calibrator accuracy is 0.02 percent. Measurement accuracy improves by a

factor of about 4 for transfer measurements (comparing input voltages with ac calibrator voltages). Means for extending this accuracy to 0.01 percent are discussed.

I. Introduction

CALIBRATED vibration transducers (accelerometeramplifier systems) are required in calibration laboratories and test facilities to determine the acceleration of vibration exciters, tables and fixtures. At very-low frequencies (VLF's), particularly below 10 Hz, the major source of

error in calibrating these transducers (both reference stan- dards and test items) has been the error in measuring the output voltage from the transducer [1], [2]. This is because

the measurement error of the few commercial voltmeters

(that measure VLF's) ranges from a few tenths of a percent

to over 1 percent, depending upon the voltage and fre- quency. Furthermore, calibration of these voltmeters is difficult since no commercial ac voltage calibrator operates

below 10 Hz. Voltage measurement errors are very

significant since their contribution to transducer calibration uncertainty is added at each level of the calibration hierarchy.

To meet present and expected future requirements of vibration and other very low frequency measurements, a portable rms digital voltmeter (DVM) was designed to measure voltages from 2 mV to 10 V over the frequency range of 0.2 Hz to 50 Hz. 1 Since the means for calibrating the voltmeter to the required accuracy below 10 Hz did not

exist, an ac voltage calibrator was also developed and

incorporated into the same instrument. This calibrator,

Manuscript received February 20, 1978. This work was supported in part by the Calibration Coordination Group of the Department of Defense. The author is with the Electricity Division, National Bureau of Standards, Washington, DC 20234.

1 The design was later modified to extend the range to 0.1 Hz.

INPUT a RMS/DC 0 tow

PASS

FILTER

M oc AMPL C0NV DVM

Fig. 1 RMS digital voltmeter used in ac voltmeter/calibrator.

covering the frequency range of 0.2 Hz to 50 Hz,2 is intended not only for calibration of the companion voltmeter but also

for the calibration of other voltmeters. In addition, the

calibrator facilitates transfer measurements. The accuracy objective for the voltmeter was 0.1 percent

of reading above 2 Hz and 5 mV and 0.25 percent of reading

for lower frequencies and voltages. As explained in the next

section, the minimum response time obtainable in rms

voltmeters is directly proportional to the period of the input

voltage. Commercial rms DVM's require 15-20 periods for a response to within ±0.1 percent of the input change. This

extrapolates to 150-200 s for 0.1 Hz. Since slow response

limits an instrument's usefulness, the design objective also was to decrease the response time below these figures as much as was practicable.

II. RMS Voltmeter Design

A. Design Considerations The rms voltmeter designed for this instrument follows the conventional approach shown in Fig. 1. In describing the operation of this circuit and the following circuits, rms and

instantaneous values will be represented by upper and lower

case letters, respectively. The ac input voltage £( is amplified

to a voltage level Ea suitable for the rms/dc converter and

converted to a unidirectional voltage Ea. The converter

transfer characteristic E0 /Eais ideally a constant, adjustable

to unity. 3 Since the input voltage to the converter ea

is sensed by a square-law device, the converter output e0 consists of a dc component £dc and a ripple component e r whose rms

value is £r ; hence,

E0 = JEl + E) ~ £dc(l + £,

2 /2£dc). (1)

If the amplifier gain is G and £„/£,, = l.then E0 = Ea = G£,. For accurate measurement of Eh the voltmeter reading

should be E0 /G; however, the dc DVM responds to £dc (since it is average-responding) and displays the value £dc /G. Therefore, it is seen from (1) that the ripple value £P causes a fractional error of £r2 /2£dc in the displayed voltage.

2 The design was later modified to extend the range to 0.1 Hz.

3 In practice, £„/£„ varies somewhat with the frequency of the input voltage This rms/dc conversion error will be discussed in later sections.

646
Thus ripple filtering is required in the converter to limit this rms/dc conversion error to an acceptable level. This constraint still allows a relatively large peak-to-peak value of er and would, if not reduced, cause intolerable fluctuations in the displayed voltage. These fluctuations, in effect, reduce the DVM accuracy. Therefore, a low-pass (LP) filter is used to reduce the peak ripple (in eM ) to an acceptable level. If the input voltage e, is a waveform with a crest factor greater than 1 (e.g., a sinewave), the time constants in the converter and LP filters must be increased in direct proportion to the period ofe, in order to maintain acceptable ripple

levels. Consequently, the minimum rms voltmeter response

time is proportional to the period of e{. For a given

frequency (and a given voltmeter design), response time can

only be decreased by decreasing the filter time constants,

resulting in increased voltmeter error (including fluctua- tions in the displayed voltage). Thus a tradeoff exists between minimum response time and minimum voltmeter

error.

B. Error Budget

The error budget of 0.1 percent above 2 Hz was allocated

as follows:

rms/dc conversion error 0.02 %

calibration errors 0.03%

temperature effects/°C 0.02%

gain-offset drift/30 days 0.03%

The error budget of 0.25 percent below 2 Hz included the above items, plus the following:

rms/dc conversion error from ripple (£r ) 0.05%

rms/dc conversion error from sensing element 0.05%

peak fluctuations in displayed voltage 0.05%

Much smaller fluctuations (0.01-0.02 percent) in the displayed voltage are desirable when using the voltmeter for ac/ac transfer measurements where the input voltage is measured by comparison with a calibrator voltage of ap- proximately the same frequency. The gain-offset error budget was not considered critical since corrections are

readily made (2 adjustments) using the voltage calibrator.

C. Detailed Design

1) Input Amplifier: The input amplifier is direct-coupled

and has gains from 0.5 to 2500, selectable in 12 ranges of 1-2-5 gain sequence. The amplifier consists of two cascaded

low-noise low-drift operational amplifiers with buffering to provide high-input impedance and power capability with low distortion at its output. The sequence of voltage ranges

limits the nominal input range over which the rms/dc

converter must operate from 2 to 5 V. Reed relays in DIP

configuration are used for range switching. The frequency

response for all gain ranges is flat from dc to 50 Hz to within

0.01 percent or better. 2) RMS/DC Converter:*

a) General description: The rms/dc converter in this instrument (Fig. 2) is a feedback type, employing a differen4 Bibliographies on rms/dc converters may be found in [3] and [4]

Fig. 2. Functional block diagram of rms/dc converter

tial multijunction thermal converter (MJTC) for the sensing

element. 5 The differential MJTC contains two matched

MJTC's connected so that the dc components of their outputs oppose each other. The outputs e L (r) and e 2 (t) have

dc components Edl and Ed2 proportional to the input power

to heater elements H x and H2 , respectively. Therefore,

Eix = K x E\lr x and Ed2 = K 2 E

2Jr2 (2) where Ea and E0 are the rms values corresponding to e a

(t

) and e0 (t), K, and K2 are constants of proportionality and r t and r

2 are the resistance values of H, and H 2

. It will be shown later that the circuit is constrained to allow only

negative feedback, i.e., the voltage fed back to H2 always

tends to decrease the difference between Edl and Ed2 . If the loop gain is very large, Edl - Ed2 ~ 0. Equating the expressions in (2), El ~ K x r 2 E

2 /K2 r, or E^y/K^IK^E,. (3) Hence, Ea

is proportional to Ea

, the rms value of the input

voltage ea(t). To simplify later discussions, it is assumed that Kj/r, = K2 /r2 = B, i.e., K x r2 /K 2 r x = 1. Equation (3) is also valid if square-rooting circuit SR is omitted and point /connected to point o. However, omitting

circuit SR causes the rms/dc converter circuit to be nonlinear, since the MJTC in the feedback loop has a nonlinear

transfer characteristic (E d2 = BE2

0 ). This nonlinearity, in turn, causes the response time of the converter to be different

for input step increases than for input step decreases [7], [8]. This effect can be removed by placing a squaring amplifier

between points/and h [8], [9], or by linearizing the con verter

circuit using circuit SR, as shown in Fig. 2. The latter method

has the advantage of maintaining constant dc loop gain,

independent of E„. b) Detailed description: Circuit details of the rms/dc

converter are shown in the upper part of Fig. 3. As explained

in the next section, resistor Rl, matched to the resistance of heater H l (~ 1 kQ), is used to reduce the low-frequency

ac/dc difference of the MJTC input section. 6 Resistor R2 is added to equalize the rms values of the input and output

voltages. Adding these resistors decreases the effective value

of B by a factor of 4. (See statement following (3) for definition of B.) Amplifier AR 1 is a low-noise chopper-stabilized amplifier

which provides high loop gain. The circuit SR (Fig. 2)

5 The MJTC is described in [$]. Differential MJTC's have been used

previously in rms/dc feedback converters operating above 50 Hz [6]. 6 The ac/dc difference is denned as 6 = (Qa - Qd )IQd where Qa and Qd are the ac and dc inputs which produce the same thermal converter output

(dc component only).

647
Fig. 3. RMS/dc converter and signal conditioner

consists of amplifier AR2 and multiplier M, which functions

as a squaring circuit in the AR2 feedback loop. The constant

k of the SR circuit's transfer characteristic (e 0 = k^/e~f ) is adjusted by the value of R3. Diode CR1 restricts the feedback voltage e0

(t) to positive values (negative feedback

only).

c) Sensing element: An MJTC was selected for the sensing element of the rms/dc converter because of its negligible ac/dc difference over a large part of the 0. 1 -50 Hz

frequency range. It will be shown later that this factor is important for the calibration of the ac calibrator. For input frequencies sufficiently high that negligible

temperature variation of the thermal converter heater occurs during a cycle (because of the heater thermal time

constant), deviation from square-law response has no effect on d and the ac/dc difference is very small. At very-low

frequencies (VLF's), the heater temperature varies with instantaneous input power and external filtering is necessary

to smooth the output voltage. Lack of square-law response now increases d. It is shown in Appendix I that the low- frequency ac/dc difference of a thermal converter (TC), caused by lack of square-law response, is

Sm = b'I

2 m2/4 (4) where / is the rms value of the input current and m is the ratio of peak to average value of the output electromotive force (EMF) (before external filtering is applied). Constant b' is a measure of the TC's deviation from a square-law response, may be of either sign, and can be determined from dc measurements (see (A. 3) and (A. 12)). Resistor Rl (Fig. 3), matched to the resistance of heater

//,, reduces 6m in two ways: 1) It minimizes the dependence

of input power on small cyclic resistance changes [10] which

decreases b\ and 2) it reduces S m by decreasing / to a

full-scale value of 2.5 mA (rated current for the MJTC is 5 mA). The value of b' should be specified so that S m < 0.1 percent for m = 1. For the MJTC used in this instrument, m

is approximately 0.6 and 0.1 at 0.1 and 1 Hz, respectively.

Therefore, 5m < 0.04 percent at 0.1 Hz and causes and

equivalent error in rms/dc conversion, which is within the

allocated error budget. At 1 Hz, <5 m < 10 ppm and decreases

to zero below 10 Hz. The ac/dc difference from other causes

(such as Thompson and Peltier effects) is less than 1 ppm up

to 10 kHz [10], [1 1], and is expected to be constant over the voltmeter's frequency range. The rms/dc converter has essentially the same ac/dc

difference as the MJTC if its output ripple is negligible

(maximum filtering used). Consequently, the ac/dc differ- ence of the rms voltmeter (input applied to the TC DIRECT

terminals) changes less than 10 ppm from 1 to 10 Hz and is negligible above 10 Hz. This voltmeter characteristic is very

useful in calibrating the ac calibrator. In the following discussions, each section of the differential MJTC is treated as a device with a single time constant, represented by the 0 63 percent step response. The MJTC

used in this instrument has time constants of approximately

1 s for each section, represented by the symbol t ,. Therefore,

the transfer characteristics for the MJTC may be written

ei (t)=Be2a (t)/(l + z iP ) (5) and

e2(t) = Bel(t)/(l + T lP) (6) where p = d/dt and B is a dimensional constant defined

earlier (see statement following (3)). d) Accuracy and response time: Since it is most difficult to meet the accuracy and response time objectives at the lowest frequencies, the converter design requirements will be

examined at 0.1 Hz. The transfer characteristics for the

circuit of Fig. 2 are derived in Appendix II-A for the critically damped and overdamped cases. These cases correspond to

circuit conditions of col = b

2 and col < b2

, where b = lfr lt coq = Bk 2 /ti RC, R is the MJTC output resistance, and C is the integrating capacitance.

The response of the critically damped circuit to an input

ea {t) = ^/2Vj sin cot is given in (A.21). The transient and

steady-state parts of that equation are

e0{t)t ^-ibte- b'V, (7) and

e0(t)ss ^{\-~-^cos2cot\Vi (8)

where

X = b

2 /{b 2 + 4co 2

) and co = Inf.

The second term of (8) represents the rms/dc conversion

error caused by ripple and is 0.12 percent at 0.1 Hz. The

response time is obtained by equating (7) to 0.0005 K, and is 18.2 s. 7 The error budget of 0.05 percent for conversion error from

ripple can be met by decreasing col (increasing C or decreas- ing k), resulting in an overdamped circuit. The response of

the overdamped circuit to ea (t) = yjl K sin t is given in

7 Unless stated otherwise, "response time" refers to the time required for the output voltage to reach its final value +0.05 percent of the voltage change.

648
(A.23) (applicable for co o < 0.95b). The transient and steadystate parts of that equation are ^*&g£%p£E3&vt ( 9)

and

y

2 Y

e0 (t)ss - 1 1 - - Jcos 2a)t (10)

where Y = a>l /[2wv

/4ft

2 - 2a>o + 4ar]. The second term of

(10) represents conversion error from ripple. 8 Equating this term to 0.0005 Vt yields to

20 = 0.61b2 at 0. 1 Hz, the value of u} 20 which will reduce the conversion error to 0.05 percent. Substituting this value for a>l into (9) and equating it to 0.0005 Vt give 34 s for the response time. Since this increase in response time was considered unacceptable, a circuit was

designed to compensate for ripple error, thus making it feasible to use the critically damped converter. This ripple error compensating circuit is part of the signal conditioner

circuit described later. 3) Low-Pass Filter: The filtering following the converter must reduce the peak ripple to 0.05 percent and should have

less than 0.05 percent transient overshoot for input voltage

steps to the rms/dc converter. In the worst case of/ = 0.1 Hz,

the ripple from the converter must be reduced by a factor of

136. These are minimum requirements (see error budget)

and can be met by 1) LP filtering, 2) integration over an

integral number of periods of ripple or 3 ) by a combination

of LP filtering and fixed period integration. Method 2) was

considered to be somewhat beyond the scope of this project. 9 Comparison of the other two methods (Appendix

II-B) was limited to filters whose poles are all real and equal.

The comparison shows that method l) has the shorter response time if the filter has 5 or more poles. However,

method 3) is more appropriate for ac/ac transfer measurements, since the fluctuations in the voltage indication are zero for frequencies of 0.05, 0.1, 0.15, Hz (includes calibrator frequencies) and are less for most other frequencies. For this reason, method 3) was used to perform the

filtering and consists of a 2 pole LP filter and an integrating

dc DVM. The filter is part of the signal conditioner circuit, described next. 4) Signal Conditioner: The signal conditioner (lower half of Fig. 3) consists of a 2-pole LP filter, a ripple error compensating circuit, and a voltage follower (AR3) with a gain of 2. The output of the rms/dc converter e 0

(t ), consisting of dc and ripple components Edc and e r

, is applied to the

signal conditioner for attenuation of <? r and correction and

amplification of £dc . The attenuation factor of the LP filter (sections R1-C2

and R8-C3) is 10 on

8 The second term of (8) and (10) corresponds to the second term of (1). 9 A long-range program is underway at NBS to develop an all-digital system for measuring and analyzing VLF waveforms in time periods approaching the theoretical minimum.

10 Attenuation factor is defined here as the reciprocal of the transfer

characteristic.

wherea= 1/(R7)(C2) = 1/(K8)(C3). As indicated in Appendix II-B2, a maximum ripple attenuation factor of 11.7 is required in the LP filter at/ =0.125 Hz (ripple frequency =

2f). This corresponds to a = 0.48. Larger a (smaller time

constants) are used for / > 0.5 Hz. The ripple error compensating circuit consists of Ql, R5,

R6, and CR2. This circuit compensates for rms/dc conversion error (from excessive ripple er ) by injecting a positive

charge on C2 during the positive half cycles of er

. The

amplitude of e r is proportional to £dc . Therefore, because of the nonlinear V — I characteristic of diodes, CR2 presents an increasing impedance to ripple as £dc decreases. This

effect is compensated by Ql which is biased so that its drain-source resistance decreases as £ dc decreases. The

compensating circuit reduces conversion error at the lowest

frequencies to well below the 0.05-peicent objective. In addition to functioning as an active filter, AR3 serves to amplify £dc to the 4 10 V level required for the dc DVM. The

overall temperature coefficient of the rms DVM is compensated to within ±0.015 percent/°C by using a Sensistor

in the feedback circuit of AR3.

5) DC Digital Voltmeter: The dc digital voltmeter consists of a voltage-to-frequency (V/F) converter and a frequency counter with digital display. The V/F converter has an input voltage range of 0 to 10 V (only the 4-10 V

range is used) and a transfer characteristic of 1000 Hz/V.

Since the number displayed by the frequency counter must

be proportional to the input voltage, the pulse rate output of the V/F converter is effectively scaled inversely to the gain

(scaling) of the input amplifier and the integration period

(either 1 or 10 s). This is accomplished by pulse rate scaling of 1, 2, and 5 (using a counter) and decimal point selection. The ripple attenuation factor (versus frequency) of the

integrating dc DVM, and its effective response time are given in Appendix II-B2.

III. Calibrator Design

The error budget for the rms DVM allocated 0.03 percent

for calibration errors. Since up to 0.01 percent calibration uncertainty results from lack of resolution in the displayed

voltage, 0.02 percent was the accuracy objective for the ac

calibrator. The voltage calibrator consists of a compact 7-digit

Kelvin-Varley (K-V) divider fed by a reference voltage of 1 V (either ac or dc) and buffered at its output by a voltage

follower with selectable gains of 1 and 10. The frequency response of the K-V divider (set to all fives) and buffer amplifier is flat to within 0.005 percent through 50 Hz. The

measured nonlinearity of the K-V divider is 0.005 percent

for a dial setting of 0020000 (2-mV output), decreasing to 0.002 percent or less at 1000000 and above. The 1-V dc

reference is derived from a precision 10-V reference supply.

The ac (sinewave) reference is shown in Fig. 4. The

sinewave is generated by a 7-bit counter-ROM-DAC combination so that 128 input pulses are required per sinewave

cycle. The necessary input frequencies are obtained using a

clock frequency of 6400 Hz and scaling it 5, 25, 50, 100, 250,

and 500 times using a series of counters. The amplifier is designed to have an output of 1-V rms and sufficient filtering to reduce the contribution of all harmonics generated by the

649
"7" Zl

CLOCK COUNTERS

T

SINE WAVE

GENERATOR

(COUNTER- ROM-DAC)

Fig. 4 Smewave reference

"W1» I

IlKT Otl o

| FUNCTI0N|

A

i ;

J V RA NGE MV J3\ V

4^-

) ' w0

I a

I PfBlOO |

RMS.'DC

| CONV &

SIG CONO

f REG COUNTER

& DISPLAY

1^ CAl fREQ ]

I | CAl VQITS ]

I | RANGE |

TO A | Q i [jo] to B .1

S El

[ CAl Quf~|

OlTA&t CAtlSRAIOH

Fig. 5. Simplified block diagram of ac voltmeter/calibrator

digital-to-analog converter (DAC) to less than 20 ppm.

Capacitances Ca , Ch

, •• are chosen so that capacitance x frequency = a constant (to within 0.2 percent). Therefore, attenuation of the fundamental and harmonic components

remains constant for all calibrator frequencies. Also, the phase shifts of the harmonics are essentially the same for all calibration frequencies. The circuit was designed to have a small output amplitude sensitivity to a change in capaci- tance or frequency and is approximately 27 ppm per percent

change.

The flatness (constancy) of the ac calibrator voltage with frequency was determined by using the ac voltmeter (input

connected to theTC DIRECT terminals) to measure 1,2, 10, and 50- Hz calibration voltages at the 5-V level. The voltage

indications for 1, 2, and 10 Hz fell within a 20 ppm range; however, each reading had an uncertainty of 20 ppm because

of lack of resolution. As described earlier, the change in the ac/dc difference of the voltmeter over this range is < 10 ppm.

Therefore, the worst case difference between the 1, 2, and

10-Hz calibration voltages is 70 ppm. The voltage indication

for 50 Hz was 40 ppm less than for 10 Hz, probably because

of the finite risetimes and small glitches between steps in the DAC output. Since the range of measured amplitudes for 1, 2, and 10 Hz was less than 70 ppm, these factors evidently

have little effect on the calibrator output in this frequency

range. More important, there is no reason to believe that these factors or any others have a more adverse effect from

0.1 to 1 Hz than in the 1-10 Hz range. Hence, it is unlikely

that the calibration voltages from 0.1 to 10 Hz fall outside a 70 ppm range. Since the ac/dc difference of the ac voltmeter

is too large to accurately compare the calibrator voltages

from 0.1 to 1 Hz, a special "viewing circuit" developed for another project [12] was used to compare the peak-to-peak

values of the calibrator voltages from 0.1 to 10 Hz. The

measured values fell within a 38-ppm range, adding

confidence to the 70-ppm range assigned above. Basing

differences between the rms values of voltages on the

differences between the peak-to-peak values appears to be

valid since all necessary precautions were taken to insure

that the wave shape is the same for all frequencies. The

calibration voltage at 50 Hz is easily trimmed (by decreasing

Fig 6 AC voltmeter/calibrator.

Ca of Fig. 4 slightly) to be the same as at 10 Hz. Thus since

the ac calibrator voltage is essentially independent of fre- quency, it is sufficient to calibrate it at only one frequency.

The ac calibrator is temperature compensated to within ± 0.005 percent/°C, using a Sensistor in series with R9 of the

amplifier. For dc voltages, the calibrator temperature

coefficient is within +0.0025 percent/°C (average of both

polarities). Calibration and overall accuracy of the calibra- tor will be discussed in a later section.

IV. Voltmeter/Calibrator Description

A. Operation

Fig. 5 is a functional block diagram of the ac

voltmeter/calibrator and Fig. 6 is a photograph of the instrument showing the front panel controls and connectors. When the FUNCTION switch is in the MEASURE

mode, the instrument functions as a voltmeter and measures

voltages applied to the AC IN terminals. The same is true in the CAL-EXT position, except that now the input voltage is also applied to the K V divider; therefore, this voltage can

be multiplied by any factor up to 10 and outputed at the CAL OUT terminals. This mode of operation permits wave

shapes and frequencies other than those provided in the

internal calibrator to be used as a reference. With

the FUNCTION switch set to either the CAL-INT AC or CAL-INT DC positions, calibration voltages determined by

the K-V divider and associated RANGE switch appear at 650
the CAL OUT terminals and are applied to the input of the

voltmeter. The calibration voltages are limited to ± 10-V dc

or 7-V ac. Selection of the calibration frequency is by the CAL FREQ switch. The dc calibrator is intended to be used

primarily as a means of calibrating the ac calibrator and is very accurate at 1 V and larger. Offsets in the direct-coupled

amplifiers of the voltmeter and calibrator limit the smallest dc voltage at which they can be used with good accuracy to

0.1 V. Measurement accuracy is determined by the accuracy of the voltmeter when it is used to measure voltages directly.

However, measurement accuracy approaches the accuracy

of the ac calibrator when the voltmeter is used as a transfer device (i.e., the voltmeter is used to compare the input

voltage with the calibrator voltage at the nearest frequency—two measurements). For example, if the in- dicated value E\ of the input voltage is dialed into the

calibrator, yielding an indicated value £'

c

, the correct value

of the input voltage is 2E\ — E'c. The FILTER FREQ (H, M, L) switch selects appropriate

filtering for different frequency ranges and the PERIOD (1, 10) switch selects the integrating period. It is shown in Appendix II-B2 that maximum filtering results in a response

time of about 37 s. With minimum filtering (including a 1-s integration period), the response time is approximately 19 s. This response was thought to be acceptable for general use; however as shown in Appendix II-C, it can be shortened by

modifying the converter circuit.

B. Calibration Use of low-drift film resistors and other components

eliminates the need for frequent calibrations of the instru- ment. However, the self-contained calibrator makes it a simple matter to recalibrate the instrument any time the accuracy is in question.

The dc calibrator serves as the basic voltage reference for the instrument. It is calibrated against laboratory standards

at the 1- and 5-V levels of the 1- and 10-V ranges,

respectively. 11 The average of both polarities is used. Since

the ac/dc difference of the MJTC (and the rms/dc converter)

is very small at 1 Hz and above ( < 10 ppm ), the ac voltmeter

(input connected to the TC DIRECT terminals)can be used

as a transfer voltmeter to compare ac and dc calibrator

voltages. Calibration of the ac calibrator is performed at 10 Hz, and consists of adjusting the ac reference so that the ac and dc voltages are equal at the 5-V level. The ac calibrator is then used for adjusting the gain and offset of the signal conditioner and the gain ranges of the input amplifier.

C. Overload Protection The input amplifier is given instantaneous protection

against peak voltages in excess of ± 16 V by diode-coupled

current sinks of 5-A capacity. Sustained overdrive blows a

fuse, disconnecting the amplifier. Applying calibrator volt- ages to the thermal converter (via the TC DIRECT ter- minals) cannot damage it. However, the voltmeter can be

used for high frequency measurements if driven at the TC

11 Trim adjustments facilitate calibration of all circuits.

DIRECT input by an external wide band amplifier. For this reason, the current sinks were also diode-coupled to point j of Fig. 3. Sustained overdrive may damage or open resistor Rl but will not damage the MJTC.

D. Performance Specifications

A brief list of the ac voltmeter/calibrator performance

specifications follows. Input signal range: 2 mV to 10 V in 12 ranges (1-2-5

sequence)

Frequency range: 0.1-50 Hz

Accuracy, percent of reading (23 ± 1°C): 0.1 percent for 0.5-50 Hz, 5 mV- 10 V ; 0.2 percent for 0. 1 -50 Hz, 2 mV- 10 V

Response time (see footnote 7): Approximately 40 s for 0.1-0.5 Hz range, decreasing to 20 s for 2 Hz and above (see Appendix II-C)

Temperature coefficient of voltmeter: ±0.015

percent/°C

Calibration voltages: 0 to 7-V ac and 0 to ± 10-V dc, with 7 place resolution Calibration frequencies: 50, 10, 2, 1, 0.5, 0.2, 0.1 Hz

Calibration accuracy (23 ± 1°C): For dc voltages in range of 1-10 V, ±0.005 percent (average of both polarities). For ac voltages, ± (0.02 percent + 0.4 jiV)

Temperature coefficient of calibrator: For dc voltages,

±0.0025/°C (average of both polarities). For ac voltages,

±0.005 percent/°C

Transfer accuracy: Voltmeter accuracy is improved by

a factor of 4.

V. Conclusions

An rms DVM has been developed for measuring VLF's

which represents an improvement of 4 to 5 times over

existing equipment with respect to response time and worst

case accuracy. Extensive circuit analysis (Appendix II) facilitated the choice and design of the circuits employed.

A self-contained calibrator, providing both ac and dc

calibration voltages, is used for 1 ) calibration of the companion DVM, 2) calibration of other very low frequency

voltmeters, and 3) very-high accuracy measurements, using

the DVM as a transfer instrument. After initial calibration against a laboratory standard, the dc calibrator serves as the basic voltage standard for the instrument. The calibration of

the ac calibrator is based on three methods : a) the transfer of the dc calibrator accuracy to the ac calibrator, using the DVM as an ac/dc transfer instrument, b) theoretical considerations, and c) peak-to-peak amplitude measurements. Theory is developed in Appendix I which relates the ac/dc

difference of a TC to its lack of square-law response and the measured output ripple. Using this theory, it is shown that the ac/dc difference of the DVM is less than 10 ppm above 1 Hz but increases rapidly as the frequency is decreased. Therefore, method a) is used from 1 Hz to 50 Hz and b)

below 1 Hz. Method c) is applicable to the entire frequency range and is used as an independent check on the other two

methods.

The ac calibrator accuracy is 0.02 percent of dialed voltage + 0.4 /iV; DVM accuracy (percent of reading) is 0.1 percent above 0.5 Hz and 5 mV and 0.2 percent below these

651
values. However, when the input voltage is measured by

comparison with a calibrator voltage of approximately the same frequency (ac/ac transfer measurement), measurement

accuracy improves by a factor of about 4. By upgrading or changing a few components, it appears

feasible to increase the ac calibrator accuracy to 50 ppm and

the accuracy of ac/ac transfer measurements to 0.01 percent

or better. The DVM has a very-high input impedance (input im- pedance of FET voltage follower) for all ranges except the 10-V range (for which it is 200 kQ.). Therefore, it is a simple matter to extend the DVM voltage range using an external,

single ratio, resistive voltage divider. Using a 1000/1 divider extends the range to 5 kV and changes the displayed voltage

from millivolts to volts. Transfer measurements (ac/ac) are made the same way as before; however, their accuracies are derated by the uncertainty of the divider ratio.

Appendix I Thermal Converter Characteristics

The transfer characteristic of a TC can be represented by a power series in I

2

E = Yi a„I", n = 2, 4, 6, • (A.l) on dc and ac for frequencies sufficiently high that temperature rise 6 does not vary during a cycle. If the frequency is so low that ^(f) follows [i(t)]

2 without significant delay or

attenuation, then

from which 12

e = £ a„i". (A.l) The temperature coefficients of electrical resistivity and

thermal conductivity, and the variation of the EMF versus 9

characteristic cause a4 to be significant. The a 6 and higher terms are usually smaller. Thus at VLF's,

e~ai2 + ab'i4 = ai

2 (l + b'i

2

). (A.3) For multijunction TC's, b'i

2 <? 1. If i = yjll sin cor, (A.3) becomes

ab I 4 e = a(I 2 - I

2 cos 2o»t) H — (3-4 cos 2a>r + cos 4cof)

(A.4) The average value of e over one cycle of / (i.e., a period

T = 2n/a>) is

1 r

7 lab'! 4

-I- —-— + cosine terms \dt

= al 2 +

lab'

I* (A.5)

With dc applied, Ed = al 2 + ab'I

4

. Therefore, for the same/,

Ea - Ed ab'I412 b'i

2 Ed al 2 + ab'I4

2

The conventional ac-dc difference 5 is defined by

8 =

/„ - /„ which equals

\E„-EA

2

(A.6)

(A.7)

(A.8)

where (5, denotes the limiting low frequency ac/dc difference caused by lack of square-law response. The constant b' can be evaluated from dc measurements.

Let a dc of rated value (/ r ) be applied totheTC. Its measured

output EMF Edr can be equated to

al 2 (l +b'I2

r ) = Edr . (A.9)

Let the applied current be reduced to I

2 j^Jl. If the TE has a

perfect square-law characteristic, the output voltage is now

Edr /2. This is numerically equal to

all ~^(\+b'I2

) = (A.10)

The actual measured voltage, however, should be equated to

Therefore, 13

V ss 2(ES - Ed ) EA2 ' (A.12)

When a low-frequency sinewave is applied to a TC, the

ratio m of peak ripple to average output EMF ranges (approximately) from 1 at very low frequencies to zero at high frequencies. Lack of square-law response has no effect on the ac/dc difference 5 when m = 0 and a maximum effect when m = 1. (Equation (A.8) represents the case of m = 1.) The effect on S for the case 0 < m < 1 is considered next. Equation (A 3) may be written as

ax

2 + ab'i4 = <?. b' 2 +

~a e > (A.13)

where e x = ai

2

. If i = J21 sin a>f and x is the TE time

constant, it can be shown that

e, = a/ 2 (l - m cos Ixpt) (A.14)

where m = + 4a>

2 x2

. Substituting (A.14) into (A.13)

and following the same procedure as in (A.5) to (A.8) yields

h

' m' - (A.15)

where Sm designates the low-frequency ac/dc difference (as a

function of m) caused by lack of square-law response.

Appendix II

A. Transfer Characteristics of RMS/DC Converter From Fig. 2

Cpef(t) =

e i (t)-e2 (t) R

(A.16)

1213 These derivations were suggested by F. L. Hermach, NBS, in a

private communication.

652
and

e0

(t) = Kj7fi). (A. 17)

Substituting (A. 17) into (6) and then (5) and (6) into (A. 16) gives

Bel{t) Bk 2 e

f

(t) + UP 1 + T lP

(A.18)

where p = d/dt, B, and k are dimensional constants, t, is the time constant for each thermal converter section, R is the output resistance of the thermoelement, and t 2 = RC. Rear- ranging (A.18)

2

1 Bk \ B

? 2 + -P + )ef(t)= e\{t). (A. 19)

Let 2b = 1/t ! and assume the system is critically damped,

i.e., 1/4t 2 = Bk2/t J t 2

. Then, if an input voltage ea{t) = Jl Vj sin cot is applied at t = 0 and the initial conditions are

zero,

14 the solution to (A. 19) is

*,(') =

k

b

2 (4w2

- (l+bt]e- b'

b

2 ]e- b' b2

(b

2 + 4w2

) 2 b

2 + 4o)2

(b

2 + 4w2

)

cos (2a>t + <p) (A.20)

where co = 2nf and <p = tan" 1 4bco/(b2 - 4a>2

). All of the

transient terms except bt exp ( - bt) can be omitted without

causing significant error in the calculated response time (see footnote 7 for definition). Also, the phase angle is not needed. Applying these simplifications to (A.20), substitut- ing in (A. 17) and using a binomial series approximation

yields 15

e0(t) a V, 1 -

16(b 2 + 4w2

) 2

- \bte12 b

2 + 4w cos 2a>t (A.21)

The second term is the rms/dc conversion error caused by

ripple.

If Bfc 2 /T, t 2 < 1/4t 2

, the circuit of Fig. 2 is overdamped

and the soluti on to (A. 19) contains transient terms in both exp ( - b + yjb2 - a*

2

, )( and exp (— b - ^b2 - cuq )', where cd

2

, = flfe 2/^ t,. If cd0

is less than about 0.95b, the exp (— b — Jb2 — a»o )f terms have little effect on the re- sponse time and can be omitted. Retaining only the largest transient term as before, the solution to (A. 19) for an input

e«(0 = \/2 V{ sin on is

o)l exp [-(b - Jb2 - (j?0

)t\ " 2v

/b2_r^(b - Vb2 - cog) Wo cos 2o»t

2cov/4b 2 - 2wg + 4o)2 (A.22)

Substituting (A.22) into (A. 17) and approximating as

before, 16

eo(0- ^ 1 -

Wo

64w2 (4b 2 - 2tog + 4oj 2

)

oj

2 exp[-(b2 -^/P^ly]

4^/b^^ib - ^^4)

a>o cos 2<x>t AwjAb2 - 2tog + 4(o 2 (A.23)

B. Ripple Filtering and Response Time

1) LP Filtering: As mentioned in the text, filtering must

reduce the ripple in e0

(t) by a factor of 136 at/ = 0.1 Hz. The

tradeoffs between number of poles and response time will be examined for LP filters whose poles are all real and equal.

If a unit voltage step is appbed to an fi-pole filter whose

transfer function is ce"/(s + af, the transient term which

largely determines the response time for at > 10 is 17

(«-l)!

(A.24)

The response time is obtained by equating this expression to 0.0005. The attenuation factor for this filter 18

is

[(4oj

2 + a2

)

1 ' 2 ]"

a" (A.25)

Using (A.24) and (A.25), and adjusting a to hold the attenuation factor constant at 136, the calculated response

times are 35, 30, 27, 26, and 25 s for n equal to 4, 5, 6, 7, and 8, respectively. Corresponding values for a are 0.385, 0.5 1, 0.62, 0.72, and 0.81. The overall response time of the rms/dc converter plus

filter will be computed for the 4 and 5 pole filters. Using the

transient term in (A. 2 1) as the forcing function, the response

of the 4 pole filter is

jl b a

4

\ \2 (5 + b)2 ' (s + a)

4

j

ba4 [(b - a)3f 3 - 6(b - a)2f 2 + 18Jb - a)f - 24]e" gl

(b - a)5

(A.26)

where b = 0.5 and a = 0.385. For the 5-pole filter, the response is

if" 1 ba 5

2(s + b)2 (s + a)5

I a

6

| (at)

6

2(5 + a)

7 r 2(6)!

(A.27)

14 For response time determinations, initial conditions can be considered zero if the input has been zero for at least 20 s. 15 This approximation is acceptable for the ranges of r and oj applicable

to response time and rms/dc conversion error (or ripple) calculations,

respectively.

16 This approximation is acceptable for the ranges of ( and <o applicable

to response time and rms/dc conversion error (or ripple) calculations,

respectively.

11 Symbol s is the Laplace transform variable

18 Reciprocal of its transfer function See footnote 10

653
where a = 0.51. The computed response times using the 4 and 5 pole filters are 38 and 34 s, respectively.

2) LP tiltenng Plus Integration: A second approach to the filtering requirement is to perform part of the filtering by

integration in thedc voltmeter. If the integration period is T„

perfect filtering of the ripple frequency (2f) occurs for input

frequencies of / = n/2Th where n = 1, 2, 3, 4, Minimum

filtering occurs for/ = (n + j)/2Tt and, depending upon the phase of the ripple relative to the start of the integration

period, approaches a minimum attenuation factor of 27t/7,.

If 7, = 10 s, its minimum value is 7.85 at/ = 0.125 Hz. (It is convenient to use integration periods of 1 and 10 s in the dc

DVM ) Using (A.21), the peak ripple from the converter at 0.125 Hz is VJ21.14. Therefore, the attenuation factor required for the LP filter is 2000/(7.85 )(2 1.74) = 11.7. According to (A. 25), a 2 pole filter with a = 0.48 meets this requirement. A good approximation to the overall response

time t f of the converter plus this filter results if we average

their time constants (b = 0.50, a = 0.48) and equate (A. 24)

to 0.0005.

1 (M9r^

e

_ o . 49(/ = a0005

2 6

(A.28)

from which t f = 26 s. The factor of $ is introduced by theSR

circuit. (See (A.21).) The integrator can start integrating at some time t

, prior

to t f without exceeding a residual transient value of 0.05

percent. Integrating the most significant transient in the

filter output over a 10-s period (starting at t = 1

1 ), taking its average and equating to 0.0005:

Vol

1

.ii + lO [cay

12 dt = 0.0005

10 12

- [(af)

3 + 3(ar)2 + 6(af) + 6]!| + 10

[.

(A.29)

Only the highest degree term for the lower integration limit needs to be retained, yielding

T( tl )^

1 10a 12 = 0.0005. (A.30)

Setting a equal to 0.49 and solving for t

t gives 21.6 s. There is no simple method of synchronizing the integrator with t v

If integration starts at r,, the ac voltmeter response time is 22 + 7, = 32 s. If it starts before t u two integration periods

are required and the response time approaches 42 seconds. Therefore, the average response time is 37 s.

C. Improved Converter Response Time For the upper decade or so of input frequencies, response

time is limited by the time constant x , of the MJTC. This can

effectively be shortened by placing a resistance r in series with capacitance C (Fig. 2). Equation (A. 16) now becomes

Cp

1 + rCp

*/(') =

R

(A.31)

Substituting (A.17)into (6) and then (5) and (6) into (A.31)

gives

CRpef (t)= (1 + rCp)

Bel(t) Bk 2 ef

(t)

(l + t lP) (1+Tjp)

(A.32)

Rearranging (A.32) and letting x 2 = RC and t 3 = rC, Bk2 * P

2 +

1 / Bk2 xA Bk2 - 1 + \p + MO

(A.33)

Let 2b = (1/t,)(1 + [Bk2 x 3 /x 2 ]), o> 0 = Bk2/x i x 2 and assume that b

2 = co2

,. Also, let ea

(t ) in (A.33) be a unit step. Then, solving for e f

(t) and substituting into (A.17) yields 19

e0 (t)- 1 -iO +bt-b2 x 3 ty (A.34)

If t 3 = 0 (i.e., r = 0), equation (A.33) reduces to (A.19). Let

ea (t) be a unit step and solve (A. 19) for e

f

(t ). Substituting this result into (A.17) gives 20

e0

(t) ~ 1 - i(l + bt)e~ (A.35)

The circuits represented by (A.34) and (A.35) have re- sponse times largely determined by the factor exp ( — bt). However, the b in (A.34) is (1 + Bk 2 x 3 /x 2 ) times larger than

the b in (A.35). Therefore, adding a resistancerin series with capacitance C (Fig. 3 ) shortens the response time by nearly a

factor of (1 + Bk 2 x i /x2 ). Let ea

(t ) = yjl Vi sin cor in (A.33). Solving for the steadystate value of eAt) and substituting into (A.17) results in 21

\ 16 2 cos 2a>t I V{ (A.36)

where Z = b

2 yjl + 4cj2 T2 /(4a> 2 + b2

). This equation

shows that adding r moves the - 3-dB point of the ripple attenuation versus ripple frequency characteristic up by at

least a factor of (1 + Bk2 x } /x 2 ), and changes theslopeof the attenuation characteristic from -12 dB/octave to —6

dB/octave. (Compare (A.36) with (8).) Acknowledgment

The author is grateful to B. A. Bell for helpful suggestions on the manuscript and to F. L. Hermach for helpful discussions on thermal converters.

References

[1] R. S. Koyanagi, "Development of a low-frequency-vibration calibra- tion system," Experimental Mechanics, vol. 15, no 11, p. 3, Nov. 1975.

[2] R S Koyanagi and J. D. Pollard, "Piezoelectric accelerometer low- frequency response by signal insertion methods," NBSIR 74-597, pp. 3^*, May 1975 (Final Report).

[3] F. L. Hermach, "ac-dc comparators for audio-frequency current and

voltage measurements of high accuracy," IEEE Trans. Instrum. Meas., vol. 1M-25, pp 489-494, Dec 1976

[4] Nonlinear Circuits Handbook Norwood, MA: Analog Devices, Inc., 1974

[5] F. J. Wilkins, T. A Deacon, and R. S. Becker, "Multijunction thermal

i9.jo.2i See footnote 15

654
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-27, NO. 3, SEPTEMBER 1978

converter," Proc. Inst. Elec. Eng., vol. 112. no. 10, Oct 1965.

[6] L. G. Cox and N. L. Kuster, "An automatic rms dc comparator," IEEE Trans. Instrum. Meas., vol. IM-23, pp. 322-325. 1974.

[7] H. Handler, "A hybrid circuit rms converter," in IEEE Int. Solid State Circuit Conf. Dig., vol. 14, p. 190, Feb. 1971.

[8] W. E. Ott. "A new technique of thermal rms measurement." IEEE J

Solid State Circuits, vol. SC-9, pp. 374-380, Dec. 1974.

[9] J. B. Folsom, "Measuring true rms ac voltages to 100 MHz," Hewlett- Packard J., voL 23, no. 7, p. 17, Mar 1972.

[10] F. L. Hermach, "An investigation of multijunction thermal converters," IEEE Trans. Instrum. Meas., vol. IM-25, pp 524-528, Dec

1976.

[11] F. J. Wilkms. "Theoretical analysis of the ac/dc transfer difference of

the NPL multijunction thermal converter over the frequency range of dcto 100 kHz," IEEE Trans. Instrum. Meas., vol. IM-21,pp. 334-340, Nov. 1972.

[12] H. K. Schoenwetter, "A high speed, low noise 18 bit digital-to-analog converter," to be published.

655
!> ON INSTRUMENTATION AND MEASI'RFMFNT, VOL. IM-27. NO. 4. r>F< FMBFH 1978

A Fast Response Low-Frequency Voltmeter

BRUCE F FIELD, mfmbfr. ieee

Abstract A sampling voltmeter implemented with a microprocessor has been developed to perform as a true rms voltmeter and waveform analyzer. The instrument measures to 0. 1 percent accur- acy the true rms voltage of approximately sinusoidal inputs at voltages from 2 mV to 10 V and frequencies from 0.1 to 120 Hz. The

major feature of the instrument is fast response time, with a total autoranging, settling, and measurement time of two signal periods

for frequencies below 10 Hz.

Manuscript received May 30. 1978: revised July 24. 1978 This work was

supported in part by the Coordination Calibration Group of the Depart- ment of Defense under CCG Project 77-1 11 The author is with the Electrical Measurement and Standards Division. National Bureau of Standards. Washington. DC 20234

I IsrKODl < HON

COMMERCIAL voltmeters presently available are generally not capable of measuring with high accuracy the

true rms value of voltages at frequencies below 10 Hz. A new

instrument has been developed and tested which is capable

of measuring the true rms voltage of approximately sinusoi- dal waveforms from 2 mV to 10 V with frequencies between

0. 1 to 1 20 Hz. An earlier ac voltmeter/calibrator, employing

a multijunction thermal converter device for rms-to-dc conversion had been developed at NBS [1] However, this new instrument is designed to have a very short settling time

as well as providing measurements of the rms value of the ac

656
field: a fast response low-frequency voltmeter

component of the signal, the frequency, and the harmonic

distortion of the signal, features which are not available on

the previous instrument. A research and development program at the National Bureau of Standards (NBS)'has recently been concerned

with developing and testing improved vibration acceler- ometers [2], [3]. The major motivation for improved acceler- ometers has been due to interest in accurately predicting the mean-time-between-failure and failure modes of equipment

subjected to mechanical vibration. Previously, the largest source of uncertainty in calibrating these accelerometers has been in the low-frequency voltage measurements. Also,

studies performed to evaluate new accelerometer designs

have produced inconclusive results because of the voltage measurement uncertainties. The present instrument was

developed to fulfill the following requirements of vibration accelerometer calibration systems:

1) frequency range—0.1 to 120 Hz fundamental +

approximately the first 10 harmonics;

2) input voltage range—2 mV to 10 V;

3) accuracy—0.1 percent of reading (2n); 4) minimal calibration in an easy-to-use portable instrument;

5) fast response time, i.e., the measurement should re- quire no more than 2 periods at the lowest frequency;

6) some measure of the harmonic distortion of the waveform should be made.

It should also be noted that in many cases vibration accelerometers produce slowly drifting dc offsets which are added to the ac component. It is awkward to manually

compensate for a changing offset thus it is necessary that the instrument provide for measuring the ac component of the

signal.

II. Instrument Design

Traditional measurement methods for ac waveforms all rely on some square-law device (thermal converter, log

amplifier, etc.) to provide a true rms indication. For this application, squaring modules based on log amplifiers do

not provide sufficient accuracies at low frequencies. Thermal

converters generally have large time constants due to the thermal mass of the heater and thermoelement. Although

improvements in thermal converter time constants can be made, increased ac ripple on the dc output results requiring

additional filtering or smoothing. Thus if optimized for low-frequency measurements, the response time of such an

instrument becomes excessively large. In addition, these methods do not provide any means for measuring the

distortion in the waveform.

The general approach used to implement this new volt- meter system is to use an analog-to-digital (A/D) converter

for digitizing the input signal. The digital values are transferred to a microcomputer for calculation of the desired

results. A simplified block diagram of the entire instrument

is shown in Fig. 1. The preamplifier is a differential-input instrumentation

amplifier with an external gain setting resistor that can be

Fig 1 A simplified block diagram of the sampling instrument

changed by the microcomputer. Four decade gain settings are available to accommodate the wide range of possible

input signals. A data acquisition system consisting of a sample-and-hold amplifier, a second instrumentation

amplifier, and an A/D converter are physically contained in one hybrid module. This second amplifier is normally wired

to provide only one of three input voltage ranges for the A/D

converter, 2.5, 5, or 10 V. However, external relays have been

added to allow the range to be set by the microcomputer,

thus a total of twelve gain settings is possible using combinations of preamp gain and A/D range. All these gain settings are necessary to obtain adequate resolution for all specified

inputs. The A/D converter is a 12-bit (11 bits plus sign bit) successive-approximation converter with a conversion time

of 12 /is. The timing and control section has two functions. The

first is a hardware clock that may be programmed by the microcomputer to output a pulse at regular intervals. This

pulse is used to trigger the A/D converter to make a

conversion, thus a series of A/D readings uniformly spaced

in time are generated irrespective of the operation of the microcomputer which may be busy with other calculations. The second function of this section is to route control signals

generated by the microcomputer to the gain-controlling

relays for the preamplifier and the A/D converter. The microcomputer used is a National Semiconductor

IMP-16. 1 This is a 16-bit word length multichip microcomputer. It was chosen mainly for the powerful arithmetic

instructions in its instruction set. All peripherals (display,

A/D converter, etc.) are treated as memory locations and the

instruction program (2048 16-bit words) is stored in ultraviolet erasable programmable read-only memories

(EPROM's). A small amount of read-write memory is provided on the microcomputer board for temporary stor- age of input values and program variables.

III. Measurement Algorithms

A combination of algorithms is used to implement the voltmeter and harmonic analysis functions. The harmonic

distortion is calculated by determining the discrete Fourier transform (DFT) coefficients using a fast Fourier transform

1 The use of this product does not constitute an endorsement by NBS:

other products may also be suitable for this application 657
IEEE TRANSACTIONS ON INSTRL MENTATION AND MEASIREMENT, VOL. 1M-27, NO 4, DFCEMBER 1978

|- Start measurement

- Find end of cycle

- Store readings

- Calculate results r Start new measurement

Fig. 2 Timing sequence of the measurement in relation to a typical sinu- soidal input

algorithm. Although the DFT coefficients can be used to calculate the rms value, serious errors result when the signal

is frequency modulated. Thus the rms value is determined

directly by numerically integrating the signal over one

period, producing a result that is unaffected by frequency

modulation. For both of these methods the period must first be accurately determined. Fig. 2 shows the measurement

sequence and timing relationships of the software and a

typical input signal.

When starting a measurement sequence on an unknown

waveform, the frequency may be as low as 0.1 Hz or as high

as 120 Hz. Sampling at a fast enough rate to capture a

possible 120-Hz signal would produce an unreasonable number of samples if a 0.1-Hz signal were present. A

preliminary decision based on the slope of the signal at the

start of the measurement is used to determine the proper sampling rate. All input voltages are scaled to roughly the same amplitude by the preamplifier and the value of the slope (the decision point is approximately equivalent to

5 Hz) determines whether a fast (8-kHz) or slow (500-Hz)

sampling rate should be used. Sampling continues, and the number of samples is counted until the sample values obtained are similar to the initial values. This provides an approximate measure 1-2-percent accuracy) of the

period. Problems that can occur when starting the measure- ment sequence at a signal peak are eliminated by delaying

slightly the start of the measurement until the peak has

passed. Advantages of this method are that the measurement

process does not need to wait for a zero crossing of the

signal; indeed, this algorithm does not require any zero

crossing. The algorithm performs well with a noisy signal,

and for low frequencies, gain changing (i.e., autoranging)

may be accomplished between measurement samples with- out disturbing the continuity of the overall measurement. A

disadvantage of this algorithm is the failure to operate with a square- wave input; however, this is outside the intended operation of the instrument. Once the end of the signal period is found, the preamplifier gain and A/D range are set to optimum values, the sample rate clock is set to 128 times the measured

frequency, and sample values spanning the next period are

stored in memory. It should be noted that the measured

frequency previously determined is of low accuracy, because

of limited resolution imposed by the calculation time

required between samples.

In order to correctly calculate the rms value of the input

signal, the period must be known to approximately the same

accuracy as is desired of the rms value. Thus a second

calculation using the stored sample values is performed to determine the period more accurately. The method chosen

to determine the period accurately must provide some way

of reducing the effect of noise. Basing any measurement or

decision on one or even a few input samples is likely to cause

a serious error. For this reason, an autocorrelation technique based on several samples is used to determine the maximum point of correlation with respect to the period

between the beginning and end of the waveform. The

algorithm chooses an initial period (an integral number of samples, N) and computes the following sum from the stored sample values (£,):

1 6 error function = £ |£, -£, + v |.

i = i The sum is calculated for different values of N until a minimum is found. However, due to the limited number of readings stored, the period is not determined with sufficient accuracy using only integral values for N. To provide more

resolution, the two smallest sums are used to linearly

interpolate to the minimum point between them. Computer simulations have shown that numerical inte- gration by the midpoint method provides good reproducibility for the value of the integral when measuring a noisy

sinusoid. Therefore, the rms value of the input is calculated

in a straightforward manner from the set of stored samples.

The sample values are squared and summed using 32-bit precision arithmetic. When the period is not an integral

number of samples, a small end correction is applied.

One feature of the instrument is an "ac only" function which removes any dc offset from the signal. In this case, the

true rms value is calculated normally by squaring and

summing the input samples. After dividing this sum by the number of samples in the period, the square of the average of

the input samples is subtracted from the sum and then the square root is taken. This produces the rms value of only the ac component.

A 64-point radix 2, fast Fourier transform algorithm is used to compute the DFT of the input signal [4], [5]. The

total execution time is typically 400 ms. For highest accuracy, this algorithm requires that 128 stored samples exactly span one signal period; therefore, after each measurement

the sample rate clock is reset to keep it "locked" to the signal

frequency. Although for some applications the measurement

of the individual coefficients is required, usually the operator wants a "figure of merit" or "how close is it to a sine wave?"

For this requirement, a total harmonic distortion (THD)

value is calculated. This consists of dividing the rms value of

the harmonic content by the total rms value. The THD of the waveform is then displayed on the front panel in percent of

the rms value of the input voltage.

658
field: a fast response low-frequency voltmeter

\BS LOW maUMYVOLTMTE*

• • •

Fig. 3. Photograph of the instrument with the cover removed The switching power supply for the digital circuitry (at left) and microcomputer are visible

TABLE I Comparison of the Sampling Instrument to an Analog Instrument Containing a Multuunction Thermal Converter

analog

instrument

(volts)

sampl ing instrument

(vol ts)

difference

(I of reading)

50 Hz 6.384 6. 386 +0.03

5.070 5. 070 0.00

3.159 3. 160 +0.03

10 Hz 3.159 3. 161 +0.06

5.072 5. 074 +0.04

IV. Performance Verification

One of the advantages of using an A/D converter with a sample-and-hold amplifier is that essentially a series of dc

voltages is being measured. Thus once the dynamic perfor- mance of the sampling circuitry is verified, calibration of the instrument can be accomplished using dc standards. Initial testing of the rms voltmeter function of the instrument was

performed with a specially constructed waveform

synthesizer, and an analog voltmeter containing a multi- junction thermal converter [1]. The waveform synthesizer

consists of a programmable digital memory feeding a

digital-to-analog converter. This synthesizer was calibrated

at dc and used as an ac source for the sampling voltmeter

and the analog voltmeter. At this time, the preamplifier had

not yet been installed so that measurements were restricted to a range of about 1 to 6 V. The calibration was performed

by applying a sinusoidal voltage (from the waveform

synthesizer) to the sampling and analog instruments and

recording the readings. Later, a dc voltage from a calibrated dc source was applied to the analog instrument and the input adjusted to obtain the same reading as obtained with

the ac. Agreement between the ac output of the synthesizer

(calculated from the dc calibration) and the analog

instrument/dc calibrator combination was within 0.02 per- cent for all the readings. Table I lists measurements taken at 10 and 50 Hz comparing the sampling and analog instru- ments. Differences between the instruments are all below the combined claimed uncertainty for the two instruments. Other tests were performed to verify the other instrument

functions; specifically, the synthesizer was used to produce a waveform with known harmonic distortion which was then measured by the voltmeter. Typical accuracy of the total harmonic distortion measurement was +0.2 percent ex- pressed as a percentage of the rms value of the input voltage.

V. Conclusions

Fig. 3 is a photograph of the instrument that has been

developed and tested. This approach which uses sampling

techniques to relate ac voltages to dc standards does not rely on fragile thermal converters or inaccurate log amplifiers, is less sensitive to temperature variations than analog designs,

659
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-27, NO 4, DECEMBER 1978

and provides rapid measurement of several signal properties. Also, the inclusion of a microcomputer in this design

produces a compact instrument, permits fast intelligent

autoranging, and simple (one button) operation.

References

[1] H. K. Schoenwetter, "An rms voltmeter/calibrator for very low

frequencies," /£££ Trans. Instrum Meas., vol IM-27, no 3, pp 259-

268, Sept 1978

[2] R. S Koyanagi, "Development of a low frequency vibration calibration system," NBS Rep 10529, Jan 1971

[3] R S Koyanagi and J D Pollard. "Piezoelectric accelerometer low- frequency response by signal insertion methods." NBS Rep 74-597. May 1975

[4] J W Cooley and J W Tukey, "An algorithm for the machine computation of complex Fourier series." Math Comp., vol. 19. pp. 297 301. Apr. 1965

[5] G. D Bergland, "A guided tour of the fast Fourier transform," /£££

Spectrum, vol. 6. pp 41-52, July 1969

660
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-25, NO. 4, DECEMBER 1976

AC-DC Comparators for Audio-Frequency Current and

Voltage Measurements of High Accuracy

FRANCIS L. HERMACH, fellow, ieee

Abstract—This paper surveys recent developments in high- accuracy ac-dc comparators, the basic standards for ac current and voltage measurements at audio and higher frequencies. These

instruments compare the unknown ac quantity with a dc reference. The ac-dc transfer characteristics of thermal converters, the most

widely used comparators, can be evaluated to about 10 ppm at audio

frequencies in national metrology laboratories. With recent de- velopments even better accuracies, and greater convenience with

automatic comparisons, are feasible.

Basic Principles

IT IS NOW FAIRLY well known that all measure- ments of ac current and voltage depend fundamentally

upon precise ac-dc comparators (ac-dc transfer standards), which either simultaneously or sequentially compare the unknown ac quantity with a known equal or nearly equal dc counterpart. Such comparators are de- signed and constructed for high precision and small stable ac-dc differences, 1 but need not have the long term stability of reading required in other instruments. They are used with basic dc standards to evaluate, for example,

calibrated ac voltage or current sources. These in turn are used to evaluate instruments such as digital voltmeters,

Manuscript received June 29, 1976. The author is with the Electricity Division. National Bureau of Standards, Washington, DC 20234.

1 The ac-dc difference is defined as h = (Qar — Qdc)/Qdc. where Qac and

Qdc are the ac and dc quantities (current or voltage) required to produce

the same response (output) of the comparator.

or ac-dc converters for such instruments. Since accuracies

of commercially available sources and instruments now

approach 0.01 percent (100 ppm), very high accuracies are required in the ac-dc transfer. The determination of the ac-dc differences of such

comparators has been, and continues to be, an important

task of national metrology laboratories. The ampere or the

volt is established with direct current or voltage in these

laboratories to a few ppm by electrodynamic or electrostatic instruments, which in effect weigh the electrical unit by balancing an electrical force against the force of gravity.

The pertinent constants of the instruments must be de- termined to the full accuracy of the measurement. While

in principle this could be done with alternating current or

voltage, the difficulties would be greater. The most re- producible or stable references are based on the Josephson

effect and the cadmium standard cell, which provide dc

standards. For these reasons ac measurements are referred

to these dc standards by ac-dc transfer standards whose

constants must simply be equal at the two frequencies. Either the rms, crest, or average value of an ac wave may

be compared with a dc value. The rms value is most often desired or required, because it governs the interchange of

electrical with other forms of energy. However one of the

others may sometimes be more suitable, as in iron-loss oe

dielectric strength testing.

One of these values cannot be deduced readily from

another. This is not academic. As accuracies increase, the

661
IEEE TRANSACTIONS ON 1NSTKUMKNTATION AND MKASl IRKMHNT. DKCKMKK.H 1976

common error of measuring a voltage or current of un- known waveform with an average- or peak-responding

instrument scaled in terms of the rms value of a sine wave

can become very serious. Because of their simplicity such

instruments are now widely used. To avoid compounding

the error they should be calibrated with corresponding

peak-to-dc or average-to-dc comparators, or with an ac

source having a crest factor or form factor (as needed)

equal to that of a sine wave to the required accuracy. Highly accurate peak and average ac-dc comparators have

been designed, but only over limited ranges to date. RMS-DC Comparators

Classical methods of ac-dc comparison are based on

electrodynamic, electrostatic, or electrothermic instru- ments, and have been described in earlier survey papers

[l]-[4]. Each of these comparators responds inherently to the rms value of the voltage or current because of its operating principle and its mechanical or thermal inertia. For example, the equation governing a conductor heated by a current i with all of the conductor at the same temperature is, at each instant, i 2 R = kd + MdO/dt, where R,

k, and M are parameters which in general depend upon the temperature rise 8. If the thermal inertia is high enough so that 8 is essentially constant during the cycle then, by

termwise integration over one cycle, the steady-state

temperature rise is simply 8a = I

2 R/k, where / is the rms

current. The temperature rise is thus a true measure of the rms

current / even if R and k are temperature dependent. This

inherent rms response holds under much more general

conditions as long as the cyclic variations of 0 are negligible.

Note that as an ac-dc comparator, 8a does not have to be an exact quadratic function of /. However, at low

frequencies the average value of the cyclically varying 8 is then not correctly a measure of /. Similar relationships hold for electrodynamic and

electrostatic instruments, but have not yet been synthesized with high accuracy by electronic means. Thermal Converters

The temperature rise of a thin straight conductor

carrying a given current is nearly independent of frequency

over a wide range because of the low reactance of the wire,

and can easily be sensed with high precision by one or more

thermocouples. The ac-dc differences of such thermal

converters (TC's) are quite small and have been studied

extensively in national laboratories over the past few

decades. TC's with appropriate electrical "readouts" have

almost entirely replaced the other forms of ac-dc comparators, which are at present more limited in accuracy by

frequency-dependent effects or in ranges of voltage and

current. A conventional simple TC (in ranges from about 2 to 500

mA), consists of a small evacuated glass bulb containing

a heater wire xk to 1 cm long, with the hot junction of a

single thermocouple thermally connected to its midpoint

by a tiny insulating bead. Thomson and Peltier heating

and cooling of the heater can affect the temperature on

direct current. 2 Even when the average of the two direc- tions is taken a small error remains. Because of thermal

inertia they do not affect it on alternating current (except

at low frequencies), leading to a frequency-independent

ac-dc difference [5], [6]. Fortunately this can be reduced

to a few ppm or less by proper design and selection of

materials. However, exact calculation is questionable, even though appropriate formulas have been developed, be- cause of possible effects of heat treatment on the ther- moelectric constants during fabrication. Many commercial TC's have lead-in wires of magnetic

alloys to match the temperature coefficient of the glass bulb. Skin effect in these leads can cause additional ac

heating. Even more serious ac-dc differences can occur in voltage measurements because of the increased ac resis- tance. They are minimized with nonmagnetic leads, which,

however, must have low thermal EMF's versus the heater

alloy and yet provide a good seal. Dielectric loss in the insulating bead generally varies with frequency and temperature, and can thus cause sig- nificant ac-dc differences. The differences are dependent

on current, frequency, and the voltage between the heater and thermocouple.

Thermoelectric errors are negligible in multijunction

thermal converters (MJTC's) in which many series-con- nected thermocouples are spread along the heater. Very

ingenious designs have been developed, some with as many

as 200 junctions and with bifilar heaters which further reduce these errors to theoretical values far below 1 ppm

[7]. They have other advantages as well, such as higher

rated output (up to 100 mV instead of 10). Investigations

have shown, however, that some constructions are subject

to dielectric loss and other errors [36]. Series resistors are used with TC's to form thermal

voltage converters (TVC's). These may be wire-wound or film-type resistors, switch-selected for convenience in obtaining multiple ranges, typically from 1 to 1000 V.

Cylindrical metal-film resistors mounted coaxially in metal

cylinders make it possible to compute the effects of re- sidual reactances to 10 MHz or more, at least at low volt- ages. Intercomparisons of adjacent ranges of such sets of TVC's provide valuable confirmation and also assurance

against other sources of ac-dc differences (which are not

likely to be the same for all ranges). Such tests can be made

to a few ppm at audio frequencies [8]. With two TC's of different current ratings (for example,

2.5 and 5 mA), a single series resistor can serve for two

voltage ranges. With only 6 resistors in a typical set, ranges from 1 to 1000 V can be obtained. If they are properly de- signed and constructed, only 7 intercomparisons are nec- essary to determine their relative ac-dc differences [9]. Similarly, shunt resistors can be used with two TC's of

different voltage ratings to make convenient thermal

2 They also cause dc reversal differences when the bead is not at the exact thermal midpoint.

662
HKRMACH: AC-DC COMPARATORS OF HIGH ACCURACY

Fig. I. Simultaneous ac-dc comparator.

current converters (TCC's), with ranges up to 20 A or more. Transformers can also be used to provide voltage ranges. Only the heater of the TC needs to be switched to the dc

reference. Excellent accuracy is possible at midaudio

frequencies, but the design of suitable transformers for wide voltage and frequency ranges is difficult. Simultaneous ac-dc comparators have been developed,

to reduce drifts from ambient and self-heating temperature changes, which can sometimes be serious. These

usually have dual circuits, as shown for one such arrange- ment in Fig. 1 [10]. One TVC is supplied by the ac voltage

and the other by a known dc voltage with the TVC outputs

connected in opposition to a null detector. Unfortunately

the input-output characteristics of TC's are not equal, (do not "track") even when equalized at one level.3 Thus for highest accuracy, a preliminary adjustment to null the detector with both converters connected to the same cur- rent or voltage is generally required. This is accomplished

with the adjustable resistor shown in Fig. 1, which forms

a voltage divider with the internal resistance of the ther- mocouple (not indicated).

Alternatively, the null-balance adjustment can be made

at a fixed dc current level, which is supplied to both heaters by a Zener-diode reference. The heater of one TC is then

switched to a decade range resistor (or an amplifier with

gain-control resistors) for the ac measurement. With

proper choice of components the resistor can be direct reading in volts when it is adjusted to obtain the same

output balance [11], [12]. Similar arrangements can be used with indirectly heated

thermistors, each consisting of a small thermistor bead in thermal contact with the heater wire. Appropriate bridge

circuits have been developed for the output balances. High

sensitivity is obtainable because of the high negative

temperature coefficients of the thermistors [13], [14]. Such simultaneous comparators, with TC's or thermistors, can provide known fixed ac reference currents or voltages for ac potentiometers (and are sometimes called "ac standard cells"). In a true potentiometer an unknown

voltage (or a definite fraction of it) is balanced against an

adjustable known voltage through an appropriate null

•' Output EMF's can differ by several percent with single junction TC's and 0.1 percent with MJTC's.

detector. For ac measurements the two voltages must be

of exactly the same frequency, magnitude and phase. Thus

the known voltage must be phase-locked to the unknown

and must be adjustable with high accuracy without af- fecting the stability of the reference. In addition, with the

usual tuned detector, only the fundamental component of the unknown is measured. For these reasons ac potenti- ometers have been used in the USA chiefly for special

measurements, when both phase and magnitude infor- mation are required.

The heater of a TC is easily burned out by overloads. This serious disadvantage can now be overcome, however,

by rapidly acting relays or electronic circuits, which are

available with some comparators [15]. This is very desirable, provided that normal changes in the components do

not affect the ac-dc difference. The continuing development of ac-dc comparators

makes standardization of accuracy classes, voltage and

frequency ranges, etc, impracticable. However, stan- dardization and agreement on terminology (including

accuracy), general requirements, and test methods is very

desirable, and has been accomplished in American National Standard C100.4 [16].

Use of Thermal Converters

For the measurement of an ac voltage, or the calibration of an ac voltmeter, the TVC is connected as shown in Fig.

2. With Vac applied, the output EMF is balanced against

a stable dc source with a null detector (not shown). The

TVC is switched to a known dc voltage, which is adjusted

to give the same output EMF. The dc voltage is reversed and readjusted. The detector null should be checked with

the ac voltage applied to guard against drifts. Then Vac =

Vdc (l + 6), where 6 is the fractional ac-dc difference of the TVC and V<j c is the average for the two directions. Alternatively, if the detector is linear, its scale can be

calibrated by adjusting the dc voltage by a small known

amount AV and noting the detector change AD. A fixed dc voltage (nearly equal to the ac voltage) can be set as a convenient reference, and the detector reading can be re- corded at each step in a sequence such as Vdc+> Vac Vdc~-

Then Vac = VdcU + o + s(Dac - Ddc )], where Dac and D<jc are the detector readings with Vac applied and the average

of the readings with the two directions of V^c applied, re- spectively, and s = A V/VAD. Errors from drifts are minimized if the steps are taken at equal time intervals. The switches and output circuit should be shielded to avoid interaction. Coaxial and shielded leads should be

used as shown, to minimize induced voltages, particularly

at the test frequency. The voltage drops in the leads be- tween the ac-dc switch and Vac and Vsc should be equal,

and errors from ground currents must be evaluated.4 Similar techniques are used for ac current measurements, but careful analysis of ground-current errors is

4 If a grounded instrument is used to measure Vdc it should be con- nected between the two switches.

663
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1976

Fig. 2. Sequential ac-dc transfer.

needed, particularly in the test of another instrument

which must itself be grounded.

A typical commercially available comparator has a

built-in ac-dc transfer switch, and an adjustable stable dc

source in series with a detector to balance the output EMF.

Most of them do not have dc reversal switches, however,

generally relying on TC's with low dc-reversal differences. This should be checked periodically at two or more heater

currents, for the error in the voltage measurement is % this difference, and can be much larger than the ac-dc differ- ence. Standard C 100.4 describes some simple yet practical

ways to detect and guard against ground-current errors, radio-frequency interference and other errors in the in- tercomparison of two current or voltage comparators.

These techniques can also be applied when ac-dc comparators are used for ac measurements or for the calibration of ac instruments. Comparators with 3- or 4-terminal in- puts (with properly designed shields and driven guards)

greatly reduce circuit-connection errors, but are not widely

used.

If the ac-dc difference of the comparator is known and

applied as a correction along with corrections for the dc

standards, and if proper techniques are used to made valid measurements, it is not at all necessary that the "accuracy

of the standard shall be ten times that of the test instru- ment" as is so often stated. The uncertainty in the ac measurement is simply the sum of the residual errors in the standards and the random error in the test (caused by both

the standard and test instrument). In this way maximum

use is made of the inherent accuracy of the ac-dc comparator and the dc standards. The application of such cor- rections to remove determined errors avoids this 10/1 noose, which often strangles echelons of laboratories with unnecessary accuracy requirements.

Automatic Comparators and Converters

All of these manual procedures are time consuming and

tedious. Automatically-switched comparators and ac-dc

converters (which generate a dc voltage accurately proportional to the input) have been developed to simplify the measurements. In some automatic comparators, essentially the same procedure is followed as in manually operated

instruments (with TC's of low dc reversal difference and

drift to simplify the ac-dc sequence). In others, the

switching is done quite rapidly (at about 1 Hz, for example)

Fig. 3. Basic TC feedback circuit.

to overcome drifts. The ac component at the switching

frequency in the detector is a measure of the difference between the ac and dc voltages, either directly or through

a feedback circuit [17]. Gain-stabilized amplifiers provide

wide voltage ranges conveniently, with minimum loading.For rms measurements many ac-dc converters are based on the feedback circuit shown in simplified form in Fig. 3

[18]. When Iac is applied, increases until the outputs

of the two matched TC's are nearly equal. If the trans- conductance of the amplifier is G and E = kl2 for each TC,

7d c = Gk (Iac

2 - 7dc2

). Then if a = 1/Gklic « 1, 7dc « Iac Vl - a « Iac . Interestingly, this is an rms instrument

with a linear scale (except near zero). To avoid tracking errors, decade input resistors, or amplifiers controlled by decade feedback resistors, can

provide appropriate voltage ranges at a fixed current. Al- ternatively a dual -heater MJTC can be used with fixed

resistors for tracking to about 0.05 percent over a range of

currents. Other types of converters have also been developed and

used in this general feedback circuit. One uses dual monolithic thermoelements, with two resistors and two

temperature-sensing transistors on a single chip [19]. In another, two indirectly heated thermistors, each with dual

heaters, are used with dual feedback so that each thermistor is always at the same temperature, to avoid tracking

errors [20]. These feedback circuits greatly reduce drifts and the

effective response time of the TC's (which is about 1 s). However the ac-dc differences of the input resistors and

amplifiers in these instruments must be small and stable. In addition, the design of the feedback amplifier is complicated by stability problems and by the double-frequency

ripple in the TC output at low frequencies.

A number of electronic techniques for rms to dc converters have been developed, based on approximating a square-law function with straight-line segments or with an appropriate transistor characteristic. Feedback circuits have made these more practical [21]. They have shorter response times and wider ranges than thermal converters.

Other AC-DC Comparators

Electrodynamic and electrostatic instruments have been

used as ac-dc comparators. The frequency range of an

electrodynamic instrument, however, is limited by the

664
HERMACH: AC-DC COMPARATORS OF HIGH ACCURACY

reactances of the coils. The range can be much greater for an electrostatic instrument, but the torque/weight ratio

is low at moderate voltages, and surface films can cause ac-dc differences that are difficult to evaluate. With both

types the double-frequency periodic variations in the ac torque can cause vibrations in parts of the moving system,

with resultant errors. It is difficult, also, to obtain a resolution of a few ppm even with light-beam pointers. For all of these reasons these instruments find their place mainly

in national metrology laboratories, where these problems

can be studied at length and overcome [22]-[24]. Putting a rectifier in the feedback loop of a suitable operational amplifier greatly reduces the effects of the

rectifier's imperfections. Refinements of this arrangement

have made average-responding ac-dc comparators and

converters of 20 ppm accuracy possible [25]. Instruments

with electronic or electromagnetic relays which compare

the crest (peak) value with a dc voltage have also been re- fined to better than 50 ppm, at least at low audio

frequencies [26], [27]. Rapid electronic sampling and

computing techniques, which measure the instantaneous

value of the wave at selected intervals, now make it possible

to compute the rms, crest and average values, and thus to determine crest- and form- factors with high accuracy as well [28], [29].

Evaluation of AC-DC Comparators

The ac-dc differences of well-designed and well-con- structed comparators are small and quite stable. In many

cases they need not be redetermined, once known, if all adjacent ranges (such as 10 and 20 V) can be compared to detect changes. The evaluation of ac-dc differences in an

absolute sense is difficult and time consuming. It has

generally been attempted only at national metrology lab- oratories, which can carefully establish and study ac-dc

transfer standards, develop the necessary precise inter- comparison apparatus, and then use this equipment to calibrate the comparators of many other laboratories. In

this way the standards receive wide use in establishing

compatability and accuracy in these measurements. This

is most economical, too, because the results of the research

are then available to all users. Such services are important

functions of national laboratories. These "absolute" ac-dc standards must be investigated

both theoretically and experimentally. This usually in- volves 1) theoretical analyses of sources of ac-dc errors, 2) design and construction of standards that minimize these

errors, 3) comparisons and other tests of these standards

to evaluate the errors, and 4) comparison with another type

of transfer standard which has also been carefully studied.The last step is very desirable to guard against errors from unknown sources, but it implies the development of different types of standards of nearly equal precision and

ac-dc accuracy. At the present time other types cannot match thermal converters in ranges or accuracy. Thus

comparisons of different kinds, constructions and ranges

of TC's, which are free of known sources of error, are vital. If these converters agree in ac-dc difference it is unlikely

that they have a common source of error, because of the

differences in their design and construction. Remarkably stable ac and dc power supplies are now

commercially available, with rms fluctuations of only a few

ppm in a minute or so. Extraordinary stabilities (0.5 ppm/h) have been achieved in recent special laboratory

designs [30], [31]. These, together with techniques which

largely cancel the effects of drifts and of small supply

variations, have made possible high precision in the in- tercomparison of TC's and TVC's.

Careful evaluations of TVC's from 1 to 1000 V and TCCs

from 1 mA to 1 A or more have been made in national

laboratories to about 10 ppm at audio frequencies, with

reduced accuracies at higher frequencies [8], [32], [33], [34]. Comparisons with electrostatic and crest-responding in- struments in these laboratories have provided valuable confirmation of the absence of large errors. At the Elec- trotechnical Laboratory in Japan an electrostatic volt- meter is planned, to provide the basic ac-dc transfer above

1 kV [35]. The development of MJTC's now makes even better accuracy possible. At NBS MJTC's of different manufacture and range have been compared, with a standard

deviation of less then 0.2 ppm. A number of sources of error which cause MJTCs to depart from the ideal have been

uncovered, and either overcome in new designs or compensated. The ac-dc difference of a group of 8 TVC's (6 to 9.V) is believed to be less than 0.3 ppm at 160 Hz, the fre- quency of an important absolute volt determination, and

0.5 ppm from 30 to 10,000 Hz [36]. An intercomparison among several national laboratories of 4 sets of TC's and TVC's is now almost finished. It should provide a better basis for international agreement

on the ac-dc transfer of current and voltage from 40 Hz to 50 kHz.

Conclusions

At the present time research and development have

provided ac-dc comparators which have the accuracy needed (about 10 ppm) to support commercial develop- ments in ac current and voltage measurements at audio

frequencies. With electronic aids, more rapid and convenient automated comparators are now feasible and are beginning to appear. There is no slackening in the importance or development of this field of measurements. A cataloging of all these developments would be neither

feasible nor digestible in a survey paper such as this. The

basic principles of a number of them have been highlighted, instead, to illustrate the present status and recent trends in high-accuracy ac measurements.

References

The following references in the open (published) liter- ature were selected because of their general nature or ex- tensive bibliographies, or because they illustrate recent

665
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1976

developments. A list of references published between 1945

and 1975 on high accuracy audio-frequency ac-dc current and voltage comparators is available on request from the

Electrical Instruments Section, National Bureau of Standards, Washington, DC 20234.

M. Iwamura and T. Yamazaki, "General review of maintenance of alternating current standards," Circular No. 147, ETL, Dec. 1957.

(In Japanese with English abstract, 81 references.)

F. L. Hermach, "Ac-dc transfer instruments for current and voltage measurements," IRE Trans, lustrum., vol. 1-7, pp. 235-240, Dec.

1958. T. B. Rozhdestvenskaya, "Electrical comparators for measurements

of current, voltage and power," JPRS 29373, 187 pp., 1965, National Technical Information Service, Springfield, VA 22161. (English

translation of a Russian Monograph with 112 references.) H. Helke, "Ac-dc converters and comparators for alternating current," Arch. Tech. Messen, no. J942-2, pp. 227-232, Oct. 1967; no. J942-3, pp. 255-260, Nov. 1967; no. J942-4, pp. 11-13, Jan. 1968. (In German, 137 references.)

[5] F. L. Hermach, "Thermal converters as ac-dc transfer standards

for current and voltage measurements at audio frequencies," J. Res. Nat. Bur. Stand., vol. 48, pp. 121-138, Feb. 1952.

[6] F. C. Widdis, "Theory of Peltier- and Thomson-effect errors in thermal ac-dc transfer devices," Proc. Inst. Elec. Eng., vol. 109, pt. C (Monograph 497 M) pp. 328-334, 1962.

[7] F. J. Wilkins, "Theoretical analysis of the ac/dc transfer difference

of the NPL multijunction thermal converter over the frequency

range DC to 100 kHz," IEEE Trans. Instrum Meas., vol. IM-21,

pp. 334-340, 1972.

F. L. Hermach and E. S. Williams, "Thermal converters for audio

frequency voltage measurements of high accuracy," IEEE Trans

Instrum. Meas., vol. IM-15, pp. 260-268, 1966.

E. S. Williams, "Thermal voltage converters and comparators for very accurate ac voltage measurements," J. Res. Nat. Bur. Stand.,

vol. 75C, pp. 145-154, 1971.

101 F. L. Hermach, J. E. Griffin, and E.S.Williams, "A system for ac- curate direct and alternating voltage measurements," IEEE Trans. Instrum. Meas., vol. IM-14, pp. 215-224, 1965. W. Rump, "On the exact absolute measurement of a-c voltages and

a potentiometer for calibrating a-c instruments," Elektrotecknik,

vol. 5, pp. 64-67, 1951 (in German).

121 J- E. Griffin and F. L. Hermach, "A differential thermocouple

voltmeter," AIEE Trans., vol. 81-1, pp. 339-345, 1962.

13) F. C. Widdis, "The indirectly heated thermistor as a precise ac-dc

transfer device," Proc. Inst. Elec. Eng., vol. 103B, pp. 693-703,

1956.

14| H. J. Schrader, "A-c current measurements with indirectly heated NTC resistors," ETZ, vol. 73A, p. 547, 1952 (in German).

15] "IR sensor protects transfer standard," Electronics, vol. 48, pp.

36-37, Oct. 16, 1975.

[16] "AC-DC transfer instruments and converters," American National Standard, C100.4, 1973. ANSI, 1430 Broadway, New York, NY

10018.

[17] L. G. Cox and N. L. Kusters, "An automatic rms/dc comparator," IEEE Trans. Instrum. Meas., vol. IM-23, pp. 322-325, 1974.

[18] R. Gilbert and J. Miller, "Thermal converter," U.S. Patent 2 857

569, 1958.

[ 19] W. E. Ott, "A new technique of thermal rms measurement," IEEE

J Solid State Circuits, vol. SC-9, pp. 374-380, 1974.

[20] P. Richman, "A new wideband true rms-dc converter," IEEE Trans. Instrum. Meas., vol. IM-16, pp. 129-134, 1967.

[21] P. Richman and W. Walker, "A new fast-computing rms/dc conversion," IEEE Trans Instrum. Meas., vol. IM-20, pp. 313-319,

1971.

|22] W. Smith and W. Clothier, "Determination of the dc-ac transfer

error of an electrostatic voltmeter," J. Inst. Elec. Eng., vol. 101, pp. 465-469, 1954.

[23| T. Yamazaki, S. Iwamoto and T. Ando, "Estimation of the ac/dc

differences of electrostatic voltage comparator," Bull ETL, vol. 32, pp. 1085-1099, 1968.

[24] G. Schuster, "A high resolution electrodynamic ac-dc power transfer instrument," IEEE Trans. Instrum. Meas., vol. IM-23, pp. 330-333,

1974.

[25] L. A. Marzetta and D. R. Flach, "Design features of a precision AC-DC converter," J. Res. Nat. Bur. Stand., vol. 73C, pp. 47-55,

1969.

[26] P. Richman, "A new peak-to-dc comparator for audio frequencies," IRE Trans. Instrum., vol. Ml, pp. 115-122, 1962.

[27) D. R. Flach and Louis A. Marzetta, "Calibration of peak ac to dc

comparators," Proc. ISA, paper no. 14.2-3-65, 1965.

[28] F. Deist and R. Kitai, "Digital transfer voltmeters," Proc. Inst. Elec. Eng., vol. 110, pp. 1887-1904, 1963.

[29] R. S. Turgel, "Digital wattmeter using a sampling method," IEEE

Trans. Instrum. Meas., vol. IM-23, pp. 337-341, 1974.

[30) H. Schoenwetter, "An ultra stable power supply for an absolute volt determination," Metrologia, vol. 10, pp. 11-15, 1974.

(31] G. Schuster, "A precision stabilized sine wave source for AC power

measurements," IEEE Trans. Instrum Meas., vol. IM-22, pp. 391-394, 1973.

(32] S. Iwamoto and H. Hirayama, "Ac/dc thermal converters of the ETL," IEEE Trans Instrum. Meas., vol. IM-23, pp. 326-329,

1974.

[33) C. H. Dix, "Electrical standards of measurement, Pt 1, dc and low frequency standards," Proc. Inst. Elec. Eng., vol. 122, no. 10R, pp. 1018-1036, 1975.

[34] O. P. Galakhova and T. B. Rozhdestvenskaya, "State of the stan- dardization of references and methods of dc and ac voltage measurements," Meas. Tech., vol. 18, pp. 1339-1341, Sept. 1975.

[35] K. Shida, "Ac-dc high voltage comparator with balance," Bull. ETL,

vol. 39, pp. 415-420, 1975 (in Japanese).

[36] F. L. Hermach and D. R. Flach, "An investigation of multijunction thermal converters," to be published.

666
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT. VOL. IM-25, NO. 4, DECEMBER 1976 Thermal Current Converters for Accurate AC Current

Measurement

EARL S. WILLIAMS

Abstract—A new 14 -range set of thermal current converters

consisting of shunted thermoelements has been constructed to measure ac-dc difference and ac current from 10 mA to 20 A at 20 Hz to 50 kHz. The ac-dc difference corrections for all ranges can

be determined relative to two ranges by a 7-step intercomparison

of certain adjacent ranges.

Introduction

THERMAL CURRENT converters (TCC's) consisting of ac shunts in parallel with thermoelements

(TE's) are widely used as ac-dc transfer instruments to determine ac-dc difference of other instruments and to measure ac currents by comparison with known dc currents. Shunt assemblies are sometimes incorporated into multirange instruments, but usually are individual sin- gle-range units. They are attached to a thermal voltage

converter (TVC) which contains a low-voltage low-current TE. Commercial multirange TVC's usually incorporate an

adjustable source of EMF and a null detector for monitoring the output of the TE.

Specific procedures and the necessary equipment have

not been provided by shunt manufacturers to enable the

user to make intercomparison tests between members of

a set of shunts. However such tests would provide several advantages. It would be sufficient for the manufacturer or a standards laboratory to calibrate only two or three

ranges. These ranges would serve as reference points for calibrating the other ranges by intercomparison. Periodic

intercomparisons could then be made by the user to detect any significant changes that might occur in any shunt in the set, and redeterminations of ac-dc difference relative to other standards would not be required as often. The new NBS set of TCC's, consisting of six shunts and

two TVC's (Fig. 1) is designed so that it can be intercompared in a minimum number of steps. The TVC's have

nominal inputs of 0.3 and 0.6 V and may be attached to any

one of the shunts by a coaxial connector. This permits each

shunt to be used for two ranges, as shown in Table I where

current ratings, shunt resistances, and TVC currents are

listed. For the purpose of calibration, intercomparisons are made between each of the six shunts or between alternate

ranges. The TVC's are used without shunts for the two

lowest ranges (10 and 20 mA), and they are compared with

the shunted ranges in two additional steps. Manuscript received June 28, 1976; revised August 12, 1976. The author is with the Electricity Division, National Bureau of Standards, Washington, DC 20234.

Thermoelements as TCC's and TVC's

TE's without shunts have been used for some years at the NBS as primary current converters. They can be very

accurate ac to dc transfer devices. The short straight heater and connecting wires have minimal reactance, and an in- sulating bead between the heater and thermocouple pre- vents any appreciable interaction between the current being measured and the readout instruments. The ther- mocouple output is rather low (7 to 12 mV), but it can be

monitored with very good precision with a modern galva- nometer or microvoltmeter.

However shunted thermoelements are probably pref- erable as TCC's in most other calibration facilities mainly

because the low current TE's used are relatively inex- pensive and easy to replace if burned out—an accident that can happen quite easily. Ampere-range TE's are more

expensive, and those with insulating beads are not ordi- narily available. However, a few sets have been made by

one manufacturer on special arrangement. TE's for 10 and 20 A usually have C-shaped heaters

(tubular except for opening on one side) which can cause an unusual effect. The ac-dc difference is very much de- pendent on the location of other nearby conductors, particularly the return conductor which is usually placed along

side of, and more or less parallel to, the TE heater. Apparently the ac current is distributed for minimum reac- tance and may therefore be concentrated close to, or away

from, the thermocouple which is attached to the top of the

heater. This nonuniform current distribution can cause an

ac-dc difference change of 0.03 percent or more. Rigid copper return conductors have therefore been added to these TE's and the input terminals are both at the same

end of the device. The conductors are also positioned for small ac-dc differences. Similar TE's with straight wire heaters (0.5 to 5 A) and

insulation have ac-dc differences of 10 ppm or less at 50

kHz.TE's for currents up to 1000 mA are available with in- sulation and vacuum bulb enclosures. They are adversely

affected by Peltier arid Thomson heating, which have been

discussed elsewhere [1], [2]. However, skin effect is the major concern in the practical use of these TE's. The

copper-coated nickel alloy widely used for passing current conductors through glass seals is sufficiently magnetic to produce a small increase in the ac impedance in the heater support stems causing additional heating with ac current but not with dc current. The effect is dependent on both

667
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT. DECEMBER 1976

Fig. 1. SetofTCC's.

TABLE I Thermal Current Converter Set

Fourtee n ranges are formed with six shunts and two TVCs (FG and FH) .

Current Shunt TVC

Range Resistance Current (amperes) (ohms) (mA)

0.01 10 (FG)

0.02 20 (FH)

0.03

15

10 0.06 20

0.1 0.2 3.33

10

20

0.3 10 0.6

1.0

20

1

2 0.3

10

20

36 0.1

10

20

1020 0.0310

20

frequency and current thus complicating stepup inter- comparison tests. A few special TE's have been made with nonmagnetic

Ni-Cr quaternary alloy for both the heater and supporting stem inside the glass bulb. This not only minimizes the skin

effect in the conductors near the heater, but also eliminates

Peltier heating at the heater ends since the supporting

conductors and the heater are of the same material. The heating due to skin effect is usually negligible at currents less than 200 mA for TE's used as current converters. However, if the TE is used as a voltage converter,

or as a TVC in parallel with a shunt, the additional impedance is added directly to the relatively low heater re- sistance. The effect is large down to at least 20 mA and

significant to 5 or 10 mA. At 2.5 mA, where the heater re- sistance (about 400 U) is high relative to that of the leads, the ac-dc difference at 50 kHz is usually less than 5 ppm.

6

The skin effect problem was overcome in these TVCs

by using 5 mA and 10 mA TE's with platinum input wires. They were made to NBS specifications by three cooperating manufacturers. Fortunately, platinum has an ex- pansion coefficient close to that of glass and makes a reli- able seal. Although the Peltier coefficient is large, the heating effect is small at these low currents. The voltage converters designated FG and FH agree with NBS stan- dards to 5 ppm or better both as voltage and current converters. The differences are equally small down to well below 20 Hz, and the corrections to a TCC are of course

quite independent of the shunt at low frequency [l]. TCC Construction, Connections, and

Grounding

The thermoelements are placed in insulating styrofoam

containers and mounted in short brass tubes. The input

is a GR874 coaxial connector and the thermocouple output

is through a two-contact connector which has very low

thermal EMF's at the contacts (Fig. 1). The 0.3-V TVC, designated "FG," is a 10-mA 30-Q TE

with no additional resistors. It also serves as a 10-mA

TCC- the lowest of the 14 ranges. The other TVC ("FH")

contains a 5-mA TE, with a heater resistance of about 90

ft, and a 33-ii resistor in series. The combination is paralleled by a second resistor of 39 U so that FH becomes a

0.6- V TVC and a 20-mA TCC.

The three high current shunts are made of bifilar strips of Ni-Cr quaternary alloy, insulated with 1-mil "Mylar"

and mounted between aluminum heat sinks (Fig. 1). The

shunt resistance material is gold plated at the ends, then

tinned and soldered to heavy copper end pieces to ensure good current distribution. The potential taps are near the

centers of the shunt strip and are brought out at right angles to the shunt by a coaxial conductor passing through one half of the heat sink. A flexible cable extends the out- put to a GR874 connector to which the TVC is attached. The outside of the coaxial conductor is connected to one

half of the shunt, and the center conductor passes through

8
WILLIAMS: THERMAL CURRENT CONVERTERS

a hole in that part of the shunt and the insulation to the

other half where it is silver soldered. The 10-20-A shunt resistor is 5 mils thick, 10 cm wide

and about 34 cm between potential taps. After being folded

into bifilar form the overall length is about 24 cm. The

resistor for the 3-6-A shunt is 2 mils thick and approximately 4 by 14 cm between potential taps. The 1-2-A shunt

is of the same material and approximately 1.8 by 25 cm.

Building the three shunts for the milliampere ranges

with available parts or materials was an awkward problem.

The required resistances were too high for bifilar strip

material and yet lower than the commercial resistors or- dinarily available. Tin-oxide resistors in parallel provided

a satisfactory solution. The 300-600-mA shunt (1 Q) contains ten 10-12 resistors, and the 100-200-mA range (3.33 Q) contains six 20-fi resistors. Parallel combinations of two

30-fi resistors and four 60-Q resistors were tried for the 30-60-mA shunt, but one 15-ft resistor was used as it had

the lower ac-dc difference. The resistors are mounted in 2-in diameter brass cylinders and connected between the outer casing and a wire which joins the center contacts of both the input and output connectors. (The input connector is out of view in Fig. 1.) Where more than one re- sistor is used they are arranged concentrically around the

center conductor. When two shunts are connected in series for intercomparison tests care should be taken to avoid capacitive currents that might flow in parallel with one shunt after flowing through the other. Therefore a shunt casing (or heat sink) should not be at the potential of the intercon- necting conductor as in Fig. 2. It is recommended that one

casing be grounded and the other connected to the high (or ungrounded) input conductor. The heat sinks for the bifilar shunts should be connected

similarly. The connections shown in Fig. 3 is recom- mended, and it should not make an appreciable difference whether one shunt is grounded or the other. The change

in ac-dc difference caused by changing the ground was

about 10 ppm at 50 kHz on the higher ranges of these

shunts. The effect was well under 10 ppm on lower ranges and at 20 kHz.

TCC Calibrations

Several procedures may be used for ac-dc difference measurements as described briefly in [3, section 6]. However, the TE comparator and the procedures described in

[3, section 7] were used for testing these TCC's. The TE

comparator affords a significant advantage where the power supplies are even slightly unstable. The precision

of these tests probably could not be obtained without it. A fraction of the output emf of one TE is balanced against the emf of the second to obtain an approximate null indi- cation on a detector. Small changes in the ratio of emfs

resulting from ac-dc differences may then be measured by

changes in the detector indication. The detector may be

a microvoltmeter, although for these tests a galvanometer was used whose scale factor (ppm/mm) was determined

for each ac-dc difference determination.

Shunt

Casings

Fig. 2. TCC's in series.

Heat Sinks

Fig. 3. TCC's in series.

TCC's (shunt and TVC combinations) are usually cali- brated and used as one instrument with one correction for each frequency of interest. However the shunt and TVC

corrections can be separated for all but the lowest ranges

if measurements are made of the corrections to the TVCs

as voltage converters. It is necessary to know these cor- rections if ranges are changed by changing the TVCs.

Separating the shunt and TVC corrections has another

advantage. Several TVCs may be used, and if one is acci- dentally burned out the shunts may be used with others.A third TVC, designated "FE," was assembled using an

ordinary 20-mA TE to demonstrate the accuracy with

which TVC corrections may be applied. Its correction is +22 ppm at 50 kHz and is ascribed to skin effect in the

heater lead -wires. Shunt tests and intercomparisons with

TVCs FH and FE are discussed below. The correction to the shunt and TVC combination (be)

is not exactly equal to the sum of the shunt correction (5s) and the TVC correction (6r). Separating these corrections

is complicated because the proportion of current carried by the TVC varies with range. Hermach has developed a simple approximate formula which is accurate enough at currents of 100 mA or more.

6c = m(5s + <5r) (1) where m = (I - lr)H, I is the total current, and It is the

current through the TVC. The multiplier m differs from

unity by only 10 percent at 100 and 200 mA, and its effect on be diminishes rapidly at higher currents. The formula

is more complicated at 30 and 60 mA and involves small

reactive components of the impedance which are extremely

difficult to measure. Corrections to these ranges are, therefore, determined by comparison with adjacent lower and higher ranges as explained below.

In the intercomparison tests, measurements are made

69
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1976

TABLE II Intercomparisons of TCC's (ppm)

TCCj TCC

2 AppliedCurrent Amperes20 kHz

D

50 kHz

D

FC FH 0.01 0 0 FC30 FH 0.02 + 1 +11 FG100 FH60 0.06 +10 +23

FC300 FH200 0.2 + 7 +20

FC1 FH600 0.6 - 7 -33

FC3 FH2 2 +12 +22 FC10 FH6 6 -24 -54

TABLE III AC-DC Difference of TCC's (ppm)

TCC AppliedCurrent Amperes

20

« measured

kHz

computed

50

measured

1 kHz

6 computed

FG 0.01 - 1 0 0 - 2

FH 0.02 0 0 - 2 - 2

FC30 0.03 + 3 + 1 +10 + 9 | FH60 0.06 + 3 0 +12 + 14

j FC1O0 0. 1 + 9 +10 + 32 + 37 1 FH20D 0.2 + 8 +10 + 36 + 37

j FG300 0, 3 +13 +17 +51 + 57 FH600 0.6 +19 +17 +53 +57

FC1 1 + 10 +10 +24 +24

FH2 2 + 12 + 10 +28 +24

FG3 3 +21 +22 +44 +46

FH6 f H8 +22 +36 +46

FG10 10 + 8 - 2 + 3 - 8

j FH20 16 + 5 - 2 + 2 - 8

of the difference between two shunt-TVC combinations.

If this difference is called "D" we may write D = da - 6 C2 (2) which becomes, after substituting from (1),

mibsi - m25s2 = D + (m26r2

~~ mrfri)- (3) At 100 and 200 mA the effect of the multiplier m ( = 0.9)

is less than 5 ppm for the corrections to these shunts and

TVC's. Since there are other uncertainties of this magnitude the effect has been neglected. However, where the

corrections are 50 ppm or more it should be taken into account. A 7-step intercomparison was made at 20 and 50 kHz,

and the relative ac-dc differences are listed under D in Table II. Measurement of ac-dc difference corrections for

all ranges were also made at the same frequencies by

comparison with NBS standard current converters most

of which are specially made TE's. These corrections are

listed under "5 measured" in Table III. In two additional columns in Table III under "<5 computed" these corrections are determined from the intercomparison tests. The base ranges serving as starting points for the computed 6's were chosen arbitrarily. The "measured" cor- rections to the 20-mA range and the 1-A range were taken

as base points and the others were computed by using (2) and the data from Table II [2]. As mentioned above, the

TABLE IV

Intercomparison of TCC's with TVC Corrections (ppm)

Appl ied 50 kHz TCCj TCC

2 Current Amperes D

6 S1 " 6 S2 FG FE 0.01 + 3 FC30 FE 0.02 + 9 FG100 FE60 0.06 +14

FC300 FE200 0.2 + 8 + 30 FG1 FE600 0.6 -52 -30

FC3 FE2 2 0 +22 FG10 FE6 6 -64 -42

( TABLE V

AC-DC Difference of TCC's with TVC Corrections (ppm)

Applied

Current

Amperes

50 kHz

TCC 4 c measured

6 S omput ed6 c computed

FC 0.01 0 0 - 5

FE 0.02 - 8 - 8 - 8

FG30 FE60 0.03

0.06

+10

+ 18 + 1 + 10

FC.100 FE2000.1

0.2 +32

+48

0 +32

+ 22 +26 +24

+24

FC300

FE600 0.3

0.6

+42

. +72

0 +42 +22 +50

+ 54 -54

FG1 FE212 +28

+53

0 +28 +22 +31

+24

+24

FC3

FE6

ib +44

+6 3

0 +44 +22 +41 +46

+46

FC10 FE2010

L6 + 3

+ ] 1 0 + 3 +22 - 9 + 4 + 4

relationship of the shunt and TVC corrections is complicated at 30 and 60 mA where the TVC's carry 33 percent

of the current. Therefore, the 20-mA TCC (FH) is used as a starting point for computing corrections to FG30 and

FG.

Similar data are listed in Tables IV and V for 50 kHz,

but TVC corrections are applied in these tables since FH

is used. The corresponding ac-dc differences for 20 kHz

were smaller, as in Tables II and III, and are not shown.

Intercomparison differences are listed under "D" in Table

IV, and TVC corrections from (3), with m \ and mi equal

to 1, are applied. (6t2 ~ 5ti = +22 ppm). The remaining

shunt differences are listed under "8sj — &S2- " At 60-mA,

equation (3) is not sufficiently exact, and, therefore, cor- rections to the 60-mA shunt and FE are not separated.

Corrections to each TCC with FE were determined rel- ative to NBS standards, and are listed under "5c" in Table

V. The corresponding values for the same shunt with FG

are taken from Table III. After applying the TVC correc- tions (5t = 0 for FG and +22 ppm for FE) the remaining

shunt corrections are listed under "5 measured."

Again, as in Table III, shunt corrections are computed

in Table V for all ranges beginning, as before, with the measured correction to the 1-A and 20-mA ranges. The

computed correction to FE60 and FG30 apply to the combination since the correction 5c cannot be reliably

separated at this level. 670
WILLIAMS: THERMAL CURRENT CONVERTERS

The 20-mA range and the 1- and 10-A ranges are rec- ommended as minimum reference points to be tested rel- ative to other standards outside the set. Intercomparison

determinations would be based on the 20-mA and 1-A

corrections, and the 10-A range would serve to verify the accuracy of the step-up tests for the hignest shunt ranges. TVC corrections would also have to be determined.

The close similarity of measured corrections for the same

shunt at different current levels shows that the ac-dc

differences are not appreciably affected by the current. This similarity in Table III also shows that the TVC's FG

and FH, which are changed when the current is changed,

have nearly the same correction. It is very unlikely that one change would offset the other on so many ranges. The

similarity of measured and computed corrections in Tables

III and V shows that the intercomparison tests can be

made quite accurately.

Using Thermal Current Converters

Ac-dc difference measurements may be made of transfer instruments and ammeters which respond nearly equally

to ac and dc currents. The TE comparator and procedures

referred to earlier may be used if the test instrument has

a thermocouple output voltage which can be connected to the comparator. Otherwise an indicator or readout in- strument is observed which is usually a part of the device

under test.

If the test instrument does not respond to direct current

it may be tested by two or three procedures, but the one

outlined below is recommended. Changes in detector in- dication are observed to determine the difference between

the ac current and an accurately measured dc current nominally equal to it. The suggested circuit is shown in Fig. 4, and the order

of currents applied to the standard TCC is indicated

(DC+, AC, DC-). The potentiometer P is adjusted for an

approximate null indication on the detector D. Detector

readings are then made for each dc current after it has been

adjusted to the nominal test value. The ac reading Ea is made after the ac has been adjusted for the desired indi- cation of the test instrument. The two readings on dc are averaged (Ed), and the ac may then be calculated:

where /dc

is the dc current, E is the TE EMF indicated by

DC+ AC DCFig. 4. Circuits for testing an ac ammeter relative to dc standards.

the potentiometer P, bs is the ac-dc difference of the standard TCC and n = A.E/&I. The factor n is a measure

of the deviation of the TE from a square-law response and

is usually 1.7 to 1.9 at rated TE current, (see [1, section

9]).

Conclusion

It has been shown that a TCC set consisting of six shunts and two TVC's can be constructed to cover fourteen ranges from 10 mA to 20 A. Intercomparison tests can be made in seven steps to assist in maintaining accurate ac-dc dif- ference corrections. Thermoelements with platinum heater

lead-wires are preferable, because the TVC corrections are

negligible. However if ordinary TE's are used the TVC

corrections can be accurately determined and applied. The

estimated accuracies from 20 Hz to 50 kHz are 50 ppm at 6 A and lower and 100 ppm at 10 and 20 A. Measurements

of ac current would require additional corrections for the dc standards used as reference and considerations of the

precision of the instrument under test. Acknowledgment

This work was supported in part by the Army Metrology

and Calibration Center, Redstone, AL. Their encourage- ment and support are gratefully acknowledged.

References

[l| F L. Hermach, "Thermal converters as ac-dc transfer standards for current and voltage measurements at audio frequencies," J. Res. Nat.

Bur. Stand., vol. 48, pp. 121-138, Feb. 1952.

[2] F. C. Widdis, "The theory of Peltier and Thomson effects in thermal ac-dc transfer devices," Proc. Inst. Elec Eng., monogr. 497M, -Jan. 1962.

[3] E. S. Williams, "Thermal voltage converters and comparators for very accurate ac voltage measurements," J. Res. Nat. Bur. Stand., vol. 75C, nos. 3 and 4, pp. 145-154, July-Dec. 1971.

|4) F. L. Hermach and E. S. Williams, "Thermal converters for audio- frequency voltage measurements of high accuracy," IEEE Trans

Instrum. Meas., vol. IM-15, pp. 260-268, Dec. 1966.

671
IEKK TRANSACTIONS ON INSTRUMENTATION AND MKASIIRKMKNT. VOL IM 25, NO 4, OKCKMMKK 1976 An Investigation of Multijunction Thermal Converters

FRANCIS L. HERMACH, fellow, ieee. and DONALD R. FLACH

Abstract—The relative ac-dc differences of a group of multijunction thermal converters (MJTC's) have been determined over

the frequency range 30 Hz-10 kHz. These MJTC's are of different ranges and were obtained from several sources. Differences were

observed at low frequencies when converters of various ranges were intercompared. For voltage measurements, the use of matched

resistors in series with the MJTC heater resistors greatly reduced

these errors and contributed to the reduction of other errors as

well. It is believed that the average ac-dc difference of this group

is less than 0.3 ppm at 160 Hz and 0.5 ppm up to 10 kHz.

Introduction

MEASUREMENTS of rms current and volt- age have been based on a technique which converts

electrical energy into heat and compares this with the heat produced by a dc signal. Single-junction thermal converters, each consisting of a wire heated by an electric current with a thermocouple to sense the temperature-rise,

have been in use for almost 3 decades at NBS for accurate rms ac-dc transfer measurements. Recently, a potentially more accurate form of the resistor-thermocouple combination, the multijunction thermal converter (MJTC), was

developed at NPL by Wilkins [l]. Verification of these MJTC's, to 0.5 ppm at 9 V and 1 59 Hz, was initially needed

at NBS for an absolute volt experiment, now well underway. Later, a goal of 1 ppm from 30 Hz to 10 kHz was es- tablished as a marked and useful advance over the present

10 ppm.

Each MJTC has a bifilar heater wire with from 50 to 200

series-connected thermocouples to sense the heater current, overcoming fundamental limitations in single-junction thermal converters [2]. Since actual converters may

depart from the ideal values, experimental comparisons

of converters of different construction, range, etc are desirable to buttress the theoretical accuracy. Unfortunately

there are at present no other suitable kinds of ac-dc

transfer standards to provide additional verification [3]. MJTC's with heaters ranging from 5 to 25 mA and with

rated output EMF's of 30 to 120 mV were obtained from

commercial manufacturers for this investigation, and

50-mA MJTC's were donated by NPL. Techniques were

developed for intercomparing MJTC's with high precision,

and a number of ac-dc errors were uncovered and over- come, to attain the desired goals.

EMF Comparator

The ac-dc difference is defined as S = (Qa - Qd)/Qd

where Qa and are the ac and dc inputs which produce

Manuscript received Juno 29, 1976 The authors ore with the Electricity Division, National Hureau of Standards, Washington, DC 2UTM.

the same output of a thermal converter. A modified Williams EMF comparator [4| was used with a commercial

chopper-type nanovoltmeter to determine A = b\ — 6 2, for two MJTC's whose heaters were connected either in series as thermal current converters (TCC's) or in parallel as thermal voltage converters (TVC's). A simplified sche- matic of the measuring circuit is shown in Fig. 1. The li-kfi three-stage divider D, (adjustable to 1 ppm) and the Lindeck potentiometer P, were first adjusted to bring the nanovoltmeter N, near zero for both positions of the key

K with dc applied, and were thereafter not disturbed

during an ac-dc determination. S was switched to ac, dc-D

(direct), dc-R (reversed), and AC in succession. In each

case the supply voltage was adjusted to bring N to zero 1

in the A position of K and the reading of N in the B posi- tion was recorded. As shown in [4], A = b\ — fa — (Nj —

A/n )/rc 2£2, where Na and Nj are the average voltage

readings with ac and the two directions of dc applied, re- spectively, and n 2 is 2 for these square-law MJTC's.

When the dc reversal difference was larger than the range of N, the reversal compensator (RC) was used. It consisted of an adjustable, 0-13-^V dc source, with negligible change on reversal. It was switched on for one direc- tion of the applied dc current and reversed for the other, so that there was no net change in the average. Carefully timed sequences of ac-dc readings, with 30-s

intervals and less than 10-ms off times were necessary to overcome serious limitations from drifts and fluctuations

in the MJTC's as well as the power supplies. In some cases Johnson noise was 0.3 ppm, and detector noise was 0.6 ppm, peak to peak, in the pass band of the detector. The

MJTC's were placed in a 0.02 m3 passive oil bath in a temperature controlled room. Even so, some had to be

discarded because of excessive drifts or large switching

transients. The detector had a 20-dB narrow-band rejection filter centered at 60 Hz, and was sensitive to common-mode

voltage at and near harmonics of its chopper frequency of 94 Hz. Significant ac outputs at twice the input frequency- were observed in MJTC's below 100 Hz (approximately

proportional to l// 1 5

), and at the input frequency above

1 kHz. Errors from these were reduced by modifying the

filter and connecting a 2-^F capacitor across the instru- ment. To verify the accuracy of the nanovoltmeter, a number

of ac-dc difference measurements were also made with a

' Close settings were not necessary for square law MJTC's. Like the prototype the potentiometer could be connected across E >, but this wus

rarely done, except to measure £•>.

- Peak-to-peak ac outputs of up to 0.5 percent, over HHX) times the normally used 0.3-«V range of the detector, were observed at an input frequency ol 30 Hz.

672
IIKKMACII AND KI.ACII: MUI.TI.H'N(TION TIIKKMAI. CONVKKTKKK

Fig. 1. Comparison circuit.

30 50 100 200 300 500 1000 FREQUENCY IN Ht

Fig. 2. Ac-dc differences of TCC's.

photocell galvanometer amplifier as a null detector, in- serting small known voltages in the detector circuit with

the reversal compensator. The results with the two detectors were in agreement to 0.2 ppm or better.

Low-Frequency Errors

Initial tests were made from 30 to 1000 Hz (a range near

5/1 about 160 Hz). Stepup comparisons as thermal current

converters (TCC's) were made by connecting, for example, two 5-mA MJTC's in parallel (outputs in series) and con- necting the pair in series with a 10-mA MJTC. The two

5-mA MJTC's were also compared, with their heaters in

series. These tests disclosed alarming differences between

ranges, roughly inversely proportional to frequency, as shown (at rated currents) in Fig. 2, with the 50 mA MJTC's

taken as a base. The cause of these differences is as yet

unknown. The error was affected by shunting the heater and was opposite in sign for TVC's. Thus it was caused by

a change in the effective resistance of the heater at low

frequencies.

As the frequency of the applied current decreases, the

heater temperature of a thermal converter begins to follow the double-frequency variation of the inst antaneous input power. This can cause low-frequency errors if the resis- tance is dependent on the temperature. However, the MJTC heaters are quaternary alloys of mainly NiCr, which

have low temperature coefficients (0 ± 20 ppm/°C). One

pair of MJTC's of the same range had nearly equal low- frequency errors but opposite temperature coefficients. The error was generally opposite in sign from that deduced

mathematically by treating all of the heater as a mass at uniform temperature, with all of the heat lost by temperature-independent lateral cooling. The thermal situation must be much more complex than this, as evidenced by the measured l// 1 - 5 variation of ac output with heater fre- quency, instead of l/f as predicted by this and other simple

models when the ratio of ac to dc outputs is small. A prototype MJTC with a NiCr heater from a third manufacturer had much larger ac dc differences; about

-100 ppm as a TCC and +120 ppm as a TVC at 30 Hz. This shows rather strikingly the necessity for experimentally verifying new designs. These errors agreed with the

sign predicted by the model, but varied roughly as l/f in- stead of l/f2 as predicted for low ac outputs.

Ac bridge measurements showed that the low-frequency

insulation resistance between the halves of a heater that had been cut in the middle was too high to account for the low-frequency error. Thus the low-frequency error is ap- parently not caused by the temperature coefficient of re- sistance of the heater or dielectric loss between the halves

of the heater. A well-known theorem states that the power in a resistor R\ fed by a constant voltage source having a fixed internal resistance R% is a maximum when R\ = /?2- From this, as shown in the appendix, the power in the heater of an MJTC used to measure voltage would to the first order be

independent of small changes in the heater resistance if a matched (equal) fixed resistor were connected in series with the heater. Analogously, for measuring current the power would be independent of small changes with a matched resistor across the heater. Since the thermocouples respond to the power dissipated in the heater, these

interesting applications of the maximum-power-transfer

theorem make possible compensated MJTC's, in which the

first-order errors from changes in effective resistance are eliminated, at the cost of doubling the current and voltage

ranges. Calculations and tests showed that matching to 10 percent is adequate for the desired accuracy. Other Errors

Other errors in voltage measurements arise from Peltier

effects at the junctions between the heater and its lead-in

wires, and skin effect in magnetic leads used in some MJTC's [1], [2]. For a given construction the errors from

Peltier effects are independent of frequency (except at low

frequencies) and current level, and are inversely proportional to the resistance, so that they could in principle be

673
IKKK TRANSACTIONS ON INSTKtJMKNTATION AND MKASl IliKMKNT, DFCKMIJKK 1 970 TABLE I TVC Comparisons

TABLE II Assigned AC DC Differences

MJTC HJTC 6 - «, - « 2 (ppm)

12 JO Hi 160 Hz tio'z B5*2 9 +0.4 -0.2- -0.2 -0.4

810*44 B5*2 9 +0.6 -0.1 0.0 -0.2

B10*2 B10*44 6 +0. 1 0.0 +0.1

9 0.0 0.0 -0.2 +0.2

eio*2 6 0.4 +0.1 +0. 1 +0.2

cio'i 6 +0.4 +0.1 +0.1 +0.6

C10*2 B10*45 6 +0.4 -0.1 -0.3 -0.1 C10*l 6 +0.3 0.0 +0.3 +0.2

B10*2 6 +0.1 +0.2 0.0 +0.4 A50'l4 6 +0.4 0.0 -0.2

A5o'l4 A50*15 a +C.2 o.p -o.: A50*15 BIO'2 9 0.0 -0.1 +0.2

NOTE: Bio'2 signifies a 10 mA HJTC , serial number 2, of manufact urer B.

B5»2 and B10*2 were mounted In unevacuated glass tubes at NBS.

A50*14 and *15 were In evacuated glass tubes with magnetic leads

All o thers were In evacuated copper containers.

evaluated by making TVC tests with different series resistors. However the low-frequency variation can be

complicated, and the evaluation is difficult in the presence

of other errors. Instead these and skin-effect errors were

eliminated by specifying MJTC's with copper leads, which

have low thermal emfs against the quaternary heater al- loys.

At higher frequencies (up to 10 kHz), tests disclosed

errors in some of the MJTC's from dielectric losses be- tween the heater and the thermocouples.3 For example,

differences of up to several ppm at 10 kHz could be ob- tained by changing the ground from one line to the other

in tests of two 5-mA TCC's in series. Dielectric errors were

also greatly reduced by the matched resistors. (See Appendix.)

TVC Tests

Since the needs were for voltage measurements, a group

of 8 MJTC's of 4 different constructions and 3 current

ranges, and with copper leads, were compensated with

matched series resistors of quaternary alloys wound on thin mica cards. 4 Each MJTC with its resistor was mounted in an aluminum box (through which oil could circulate), with

input binding posts and a 2-pin output connector of low

thermal EMF's. They were compared as TVC's at 6 to 9 V,

at 30, 160, 1000, and 10 000 Hz. The results are shown in Table I. At the important frequency of 160 Hz the results are shown more vividly in Fig. 3. The circles identify the MJTC's and the number adjacent to each arrow gives bh — 5( in ppm, where <5j, and <5, are, respectively, the ac-dc

1 Dielectric losses in the oil were negligible.

4 Two MJTC's, with 50-mA ranges, have magnetic leuds and were not used above 1 kHz. A series resistor of 200 i! was used with each, to give

a 10- V range. Tests with other resistors showed that resistance-dependent low frequency errors of the,e MJTC's were less than O.fi ppm. Series

resistors for the other MJT( "s were within It) percent of the heater re- sistances.

MJTC

6 (ppm)

30 Hz 16" Hz 1 kHz 10 kHz B10*44 n.o -0.1 -0.1 -0.2 310*45 0.0 +0.1 -0.

0.0 cio'i +0.3 +0.1 0.0 +0.3

C10"2 +0.4 0.0 -0.2 0.0 B10*2 0.0 -0.1 -0.1 -0.2

B5*2 -0.5 +0.1 0.0 +0.1 A50*]4 0.0 0.0 +0.2 A50*15 -0.1 -0.1 +0.2

i

Fig. 3. TVC's at 160 Hz.

differences of the MJTC's at the head and tail of the arrow. In the absence of systematic errors the arrows around the loops should sum to zero within the limits imposed by the random errors of the comparisons. Alternatively failure to close properly is an indication of systematic errors. Techniques described by Youden [5] can be used to eval- uate networks such as this. The values of <5 assigned in this way to each MJTC at 160 Hz are shown in each circle, on the assumption that the average of all is zero. Each value is less than 0.2 ppm, and

the closure errors are very small. Similar analyses were carried out for the measurements

at other frequencies. The assigned ac-dc differences are shown in the columns marked 5 t in Table II. The dis- crepancies are somewhat larger than at 160 Hz but there

is no evidence of systematic error, or of significant residual low-frequency errors in these compensated MJTC's.

The standard error s (standard deviation of the average

of 4 determinations) in a comparison of 2 MJTC's was 0.12 ppm. Because of the network of comparisons the standard

deviation of an MJTC with respect to the average should

be somewhat less than this.

Series Resistors

The high-quality wire-wound series resistors used with

these MJTC's have excellent ac-dc characteristics at audio

frequencies. The phase angles are small and cause only a 674
HKKMAC'H AND Kl.ACH: Ml'l.TMUNCTION THKKMAL CONVKKTENS

Fig. 4. Reversing bridge.

second-order effect on impedance magnitude. However

there is a possibility of dielectric loss in the insulation at the higher audio frequencies. (The maximum powertransfer theorem does not help here.) To determine this, the 1/1 inductive-voltage-divider bridge shown in Fig. 4 was set up to compare two nearly equal resistors R\ and Ro- A balance was made for each switch position of this "re- versing bridge" by adjusting m and n with the detector lead

d, as shown, after adjusting the Wagner arms with d

grounded. Most of the bridge errors were eliminated in this way, leaving as the difference equations, (R2 — Ri)/Ri =

2(na - rib) and a2 - a\ = y(ma ~ rrib) where/? and a are the resistive components and the phase angle, respectively, of the parallel equivalent circuit for each resistor, 7 =

UC3R, and the subscripts d and b refer to the switch positions shown. These simple equations hold to 0.1 ppm if the phase angles of C3 and m are each less than 0.1 mrad and

(0:2 — «i) is less than 1 mrad.5 A 1-kfi resistor of the same type as those used with the

MJTC's was mounted in a metal box, for four-terminalpair ac-dc measurements on Cutkosky's bridges [6] from

160 to 16 000 Hz. This served as a standard for evaluating

two 1-kQ unmounted resistors. These were paralleled to evaluate a 5-section 500-fi card. The 5 sections were in- tercompared and then used to evaluate the 300, 400, and

500-Q cards for the MJTC's. The 1300-1] card for the 5-mA

MJTC was compared with the 1-kfl card in series with 3

100-fl sections. An upper limit of 10 kHz was imposed by the effects of lead inductance and of stray capacitance from the cards

to the grounded aluminum boxes in which they were tested and used. Below this, changes in the parallel resistances were less than 0.1 ppm for drastic changes in the positions

of the resistors. The cases of the inductive voltage dividers and the shields of all leads of the reversing bridge were

grounded. C3 was a low-loss 3-terminal air capacitor, and

D was a tuned null detector. All bridge components were

commercially available. Measurements were made at 0.1, 1, and 10 kHz with 8 V applied. The results are shown in Table III as b e and a, where & c = (Rf — Rp )/Rp and a is the phase angle. Rf and

Rp are the parallel-equivalent-circuit resistances at the frequency / and the base frequency (0.1 kHz), respectively. For each resistor, a was closely proportional to /, and so is not shown at 1 kHz in the table.

5 The dividers were rated up to 10 kHz. Only the lower dials were changed on reversal, with only moderate accuracy required.

TABLE III Resistance Measurements

R

1 kHz 10 kHz MJTC (1 6 e

6 ea 6 r6

c B10*44 500 -0.1 -0.7 -0.55 -0.5 +0 .3 B10*45 +0.1 +0.3 -0.41 +0.1 -0 .1 C10*l 300 0.0 +0.5 -0.60 +0.1 +0 .2 C10

#

2 0.0 0.0 -0.41 -0.1 +0 .1 B10*2 400 0.0 +0.4 -0.47 +0.1 -0 .3 B5*2 1300 0.0 0.0 -0.26 0.0 +0 1

NOTE: All values of 6 in ppm , a In mrad. See text

.

The effect of these small errors on the magnitude of the current through the MJTC is approximately 6r = 5 e /2 —

3a2/8. This was negligible at 1 kHz and, except for one

resistor, even at 10 kHz. The average for all of the resistors was less than 0.1 ppm at each frequency, so that no cor- rections need be applied to Table II. The ac-dc differences of a TVC is 5y = 5r + 6C [2] where

5C is the ac-dc difference of the thermal converter. Since

8 V and 8 r were evaluated as indicated above, 5 C could be

determined for each MJTC. As shown in Table III, each was 0.3 ppm or less at 10-kHz.

Conclusions

This work reaffirms the value of both experimental and

theoretical investigations to evaluate ac-dc differences in an absolute sense, and to determine how well Actual ac-dc

transfer standards meet the theoretical ideals. A number

of sources of ac-dc differences were uncovered, and were

eliminated by changes in design, or were compensated for. The agreement between a group of 8 MJTCs which met

these requirements was excellent. Because of this, because

of the significant differences in construction and ranges

of these MJTC's, and because of our long investigation, we

believe it is very unlikely that the average ac-dc difference of this group exceeds 0.3 ppm at 160 Hz or 0.5 ppm at 30

and 10 000 Hz. This belief is supported by an analysis

based on estimates of the limits of error from such factors as mismatch of series resistors, Peltier-heat exchanges,

ac-dc differences of resistors, systematic errors in the comparator, and random errors. For the latter we took as the 95 percent confidence limit 2s/v7i, where n is the number of MJTC's in the group. A few measurements over several months indicate a

repeatability of 0.3 ppm or better. The long-term stability of ac-dc difference is as yet unknown, but there is no rea- son to expect drifts greater than this. Extensions to higher voltages and frequencies seem

feasible. Small ac-dc differences in the effective heater

resistance could be compensated with combined shunt and

series resistors, as indicated in the Appendix. A wide range

of film resistors could be mounted in a geometry that would allow the reactance to be calculated, so that the powerful comparison technique of [2] could then be applied.

675
IKKK TRANSACTIONS ON lNSTHUMKNTATION AND MKASl 'KKMKNT, DKCKMHKH 1976

Appendix

Compensation for Chances in Resistance

When a resistor: mR is in series with MJTC heater, R(l + a), a « 1, the current (with no reactance) is

(m + 1)R(1 + a/(m + 1))

(m + 1)R\ m+ 1/

neglecting higher order terms in a. The power in the heater

is P-TOl + a)*- y^—./i --**- + a ). (m+l)2R\ to+1 / For a fixed V and m, the power is independent of a if m

= 1 (neglecting higher order terms). Thus since the ther- mocouples respond to the power dissipated in the heater, a small difference between the effective dc and ac heater

resistances will cause no first-order ac-dc difference in a TVC with a matched series resistor. Matching to 10 per- cent will reduce a 2 ppm ac-dc difference (from this cause)

to 0.1 ppm.

Similarly for a TCC with mR in parallel with R(l + a) the power in the heater is independent of a if to = 1, so that there will be no first-order ac-dc difference with a matched

shunt. This compensation will be effective, however, only if the heater is a two-terminal resistor in the measuring circuit, and if the thermocouples are insensitive to any change in the distribution of the power. For example, an error would

occur with dielectric leakage from the heater to the thermocouple of an MJTC unless the thermocouple. output and

one end of the heater were connected together, and the thermocouple resistance was much less than the insulation resistance. Matching provides only a single compensated voltage

or current range. However, for higher voltage ranges a similar analysis shows that if a resistor nR is connected

across the heater and mR in series with the combination,

compensation is achieved if n = m/(m — 1). Acknowledgment

The authors are very grateful to F. Wilkins for the NPL

MJTC's, to R. Cutkosky for the ac-dc measurements of the mounted resistor, and to J. Sutcliffe and I. Malcolm of Guildline Instruments Ltd. for their extended cooperation. References

[1) F. J. Wilkins, "Theoretical analysis of the ac/dc transfer differences

of the NPL thermal converter over the frequency range dc—100 kHz," IEEE Trans, lnstrum. Meas., vol. IM-21, pp. 334-340,

1972.

[2] F. L. Hermach and E. S. Williams, "Thermal converters for audio- frequency voltage measurements of high-accuracy," IEEE Trans, lnstrum. Meas., vol. IM-15, pp. 260-268, 1966.

[3| F. L. Hermach, "Ac-dc comparators for audio-frequency current and

voltage measurements," to be published.

[4] E. S. Williams, "Thermal voltage converters and comparators for very- accurate ac voltage measurements," J. Res. Nat. Bur. Stand., 75C,

pp. 145-154, July-Dec. 1971.

[5] W. J. Youden, "Measurement agreement comparisons," in Proc. 1962 Standards Lab. Conf., NBS Misc. PubL 248, pp. 147-152, Aug.

1963.

(6] R. Cutkosky, "New NBS measurements of the absolute Farad and Ohm," IEEE Trans. lnstrum. Meas., vol. IM-23, pp. 305-309,

1974.

676
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-23, NO. 4, DECEMBER 1974 A Low-Temperature Direct-Current Comparator Bridge

D. B. SULLIVAN and RONALD F. DZIUBA, member, ieee

Abstract—The application of superconducting direct-current comparators to the measurement of resistance ratios is described. One

comparator consists of a binary set of ratios between 1 : 1 and 160:

1

providing for self-calibration by a buildup procedure. A second

comparator exhibiting discrete ratios of 1:1, 10:1, and 100: 1 is also described. Ratio uncertainty of less than 1 part in 109 is achieved by enclosing the ratio windings in overlapping toroidal superconducting shields. Superconducting quantum interference devices (SQUID's) serve as flux sensors for the comparators. One of these

current comparators is used to calibrate a 100-Q: 1-1) resistive divider, which at a current of 10 mA exhibits a self-heating error of 0.0023 ppm.

INTRODUCTION

RECENT papers [l]-[3] have described the application of superconducting shielding and superconducting quantum interference devices (SQUID's) to the problem of current-ratio measurements. At low current

these direct-current comparators outperform their con- ventional room temperature counterparts in accuracy and sensitivity by several orders of magnitude. This

paper describes a further advance in the design of such

comparators and the application of a direct-current comparator to the calibration of a 100:1 resistive divider and the evaluation of the resistor power coefficients [4]. The original work on low-temperature comparators was done by Harvey [1]. His concept can be readily

understood by considering a long superconducting tube

enclosing several wires. Current through one wire of the

cable results in an equal and opposite current on the

inner surface of the tube. This current returns on itself via the external surface of the tube, and one thus obtains the ideal condition wherein the same current flowing

through different windings generates identical fields outside the cable. That is, the fluxes sensed outside the tube are only the mutual fluxes of the wires in the tube

and not the leakage fluxes. To form a comparator then, one need only fashion a coil out of such a cable and inter- connect a number of windings to arrive at the desired

ratio. The null flux is sensed within this coil and care must be taken to shield the cable ends (which deviate from ideal behavior) from the null sensor. Harvey's comparator offers an elegant means of constructing a current comparator, but the application of the concept becomes quite tedious for large ratio (»10)

devices since one must essentially build up the large ratio

Manuscript received July 3, 1974. This work is a contribution of the National Bureau of Standards and is not subject to copyright. D. B. Sullivan is with the Cryogenics Division, National Bureau

of Standards, Boulder, Colo. 80302. R. F. Dziuba is with the Electricity Division, National Bureau of Standards, Washington, D. C. 20234.

by connecting together a series of unit windings. For

this reason a modified version of the shielding was de- veloped and is called an overlapped-tube shield [3]. This

shielding permits continuous windings and thus circumvents the aforementioned drawback.

Two comparators of this design were constructed. The

first (binary) comparator consists of a binary set of windings which provides all integer ratios between 1 : 1 and

160:1 and as a consequence permits a completely consistent check of ratio uncertainty. This comparator has a

sensitivity of 0.032 nA (for a 14-turn unit winding) with no ratio errors detected above the noise level (less than

5 parts in 1010 ). The second (decade) comparator contains

four windings providing discrete ratios of 1:1, 10:1,

100:1, and an additional unit winding for checking the balance of the comparator. This comparator is designed

to couple indirectly to a SQUID sensor via a flux trans- former. It exhibits a resolution of 0.06 nA for a 20-turn

unit winding and ratio uncertainties of less than 5 parts

in 109. A 100-fi:l-S2 resistive divider was calibrated using the binary comparator. The resistors were fabricated from a

silicon copper alloy [5], with the 1-ft resistor exhibiting a power coefficient of resistance as small as 0.023 ppm/mW.

At an operating level of 1 V for the 100: 1 divider, the self- heating error is 0.0023 ppm. The power coefficient of the resistors was measured over the temperature range

of 1.5-4 K, and it would appear that reasonable operation could be expected at 4 K as well as just below the A point

(2.17 K, the temperature below which the liquid helium

behaves as a superfluid).

CURRENT COMPARATOR

The shield configuration for the comparator is shown

in Fig. 1. The superconducting shield is insulated at the overlap so that it appears like a snake swallowing its

tail. The leads to the ratio windings are enclosed in super- conducting tubes with an added tube for every turn of overlap. These shields for the leads are also insulated from one another. Care is exercised to prevent any con- tinuous superconducting connection around the toroidal

shield as this would short the windings. The null flux

is sensed in the center window of the toroid either directly

with one of Zimmerman's multihole SQUID's [6] or

indirectly by a similar SQUID via a flux transformer. The performance of this type of shield is readily understood as a simple extension of the principle of Harvey's

shield. The external flux generated by every winding

within the shield results from the induced flow of a surface current on the shield. The purpose of the overlap is to reduce the nonideal effects at the ends of the tube. The

677
SULLIVAN AND DZ1UBA: COMPARATOR BRIDGK

" Leads to Coil within Superconducting Shield

Superconducting Shield

Fig. 1. Overlapped-tube shield for the comparator. The supercon- ducting shield is formed over the windings as a continuous tube which overlaps itself. The overlap is insulated and runs approximately 1| turns around the toroid. At each intersection of the

toroidal shield with the leads from the coil, a superconducting

lead shield is attached to the shield. These two tubes are also insulated from one another.

overlap acts as a simple flux attenuator that reduces

direct leakage of flux from the windings to the exterior regions of the shield. With this in mind, the required

overlap can be roughly estimated. The pertinent at- tenuation parameters include the length, thickness, and

perimeter (width) of the overlap region. As a practical matter the minimum thickness of overlap is limited by

available insulation materials. Thus, the only really

adjustable parameters are the length and perimeter of the overlap. As a conservative estimate of the attenuation, a flux reduction of a factor of 10 is assumed for every unit of the length-perimeter ratio. The shield is designed with If turns of overlap which corresponds to a length-perimeter ratio of ~9 (length = 17.5 cm and

perimeter = 2.0 cm). Thus if the windings are initially

matched to 10 percent, the 109 attenuation of leakage

flux would result in a balance error of 10-10 or less. Several methods were used to fabricate the winding

and toroidal overlapped shields. For the binary comparator, the coil, which consists of 10 sets of windings of 0.074-mm-diameter niobium wire (insulated), is wound

on a thin (0.5-mm-thick) aluminum form and impregnated with epoxy during the winding process. After the epoxy sets, the aluminum form is removed chemically

with sodium hydrozide. This results in a rigid coil, which

makes the shield fabrication much simpler. The ratio windings have relative winding values of 1, 1, 1, 2, 4, 8, 16, 32, 32, and 64. Each unit winding consists of 14 turns

of the niobium wire. The construction of the decade

comparator is similar. The prime difference is that a removable Teflon form was used in the casting of the

rigid coil. Also, rather than epoxy, alpha cyanoacrylate

adhesive was used to bond the coil together.

Both shields are composed of a lead-tin alloy (0.125

mm thick) which is formed around the rigid coil and

sealed with Rose's metal (50-percent Bi-25-percent

Pb-25-percent Sn, Tc « 8.5 K). The shield is fabricated

in roughly one-turn sections and each section is then

insulated with Teflon tape (except at one end) before the subsequent shield section is formed on top of it. As each

turn of the shield is added it is connected in series with

the underlying shield to form the continuous overlapped

Fig. 2. Binary current comparator. The can on the left, when

attached to the comparator, provides a complete superconducting

shield of the SQUID and windings from external fields. The

shielded leads from the coil leave the interior of the shield can

via another flux attenuation tube. The individual coil leads are spot welded to the niobium posts on the terminal board. The SQUID is within the toroidal shield and is enclosed by a thin brass shield which serves as a low-pass filter to isolate the ratio windings and the SQUID at radio frequencies.

tube. The leads into the coil are shielded with lead tubes

that are soldered to the main shield tube. At each point

where the toroidal shield intersects the leads to the coil, a

lead tube is connected to the intersecting portion of the

shield. These lead tubes are insulated from one another. The completed binary comparator is shown in Fig. 2. The SQUID sensor is inside of a thin-walled (0.10-mmthick) brass shield within the center of the toroid. The

brass shield acts as a low-pass filter to isolate the SQUID

and ratio windings at radio frequencies. The ratio windings and sensor are housed within a superconducting

shield fabricated from brass and electroplated with ~0.05

mm of lead. The shielded leads from the ratio windings are routed through a field attenuation tube and the wires

are spot welded to niobium posts on the terminal board. Niobium wire is used to interconnect coils to achieve the

desired ratio. The SQUID consists of a superconducting loop (or loops) of low inductance which is closed by a weak link or Josephson junction. The SQUID RF impedance is a

periodic function of the flux linking the ring with a period

of one flux quantum, ^ « 2.07 X 10"15 Wb. The flux quantum is thus a natural unit with which to describe the comparator performance. For this particular device a resolution of 10-Vo is achieved. For further details of SQUID operation the reader might consult a paper by

Zimmerman et al. [7]. The sensitivity of one unit winding of the binary comparator is found to be 3.2 X 10-7 A/V0 - At the flux resolu- tion of 10~Vu this results in a sensitivity of 3.2 X 10~" A

for a unit winding. Each unit of winding consists of 14 turns so that the sensitivity can also be expressed as ~0.45 nAt. The ratio uncertainty of the binary comparator was

678
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1974

determined by comparing windings in series opposition

in a buildup fashion. The unit windings are intercompared;

the sum of two of the unit windings is compared to the

two-unit winding; the sum of the two-unit winding and

two of the unit windings is compared to the four-unit winding; the buildup is continued in this fashion until all windings are intercompared. A fixed current of 70 mA

was passed through such matched sets of windings con- nected in series opposition, and the departure from null was then used as a measure of the ratio uncertainty. For

every measurement no error could be detected down to a

level of 10-

Vo. For the 1 : 1 comparison this results in an

uncertainty of less than 5 X 10-10 and for the (32 +

32) :64 comparison an uncertainty of less than 7.5 X 10-12

is indicated. It is interesting to note that the largest error

is expected for the last intercomparison because the

64-unit winding and the two 32-unit windings are further separated within the coil than are any of the other inter- compared sets. Thus, while a limit of 5 X 10-10 is set experimentally for the intercomparison of the unit windings, the limit is more likely to be smaller than 7.5 X 10-12

as found for the 64: (32 + 32) comparison. Since no

actual uncertainties were measured, it is assumed as a "worst case" that all errors are of the same sign and have

a value equal to the limit of error. For a ratio of 160:1

this yields an estimated error of approximately 1 X 10-9. The shielded coil of the decade comparator is 2.0 cm

long with inside and outside diameters of 1.8 cm and

2.2 cm, respectively. As with the binary comparator, it

is housed in a superconducting can with a brass shield within the window of the toroid. A one-turn coil of niobium

tape of thickness 0.05 mm and a diameter of 1.6 cm is located within the brass shield. This coil is connected

to a niobium coil closely coupled to a SQUID sensor, thus forming a flux transformer. Operation with the flux trans- former reduces the sensitivity of the comparator by

approximately 20 percent, but it eliminates the need to permanently house the SQUID sensor in the window of the toroid. Alternatively, other types of SQUID's can be

used for operation of the comparator. For this particular

arrangement a sensitivity of 0.06 nA for a 20-turn unit winding or 1.2 nAt is achieved. The unit windings were intercompared with a test current of 150 mA by the technique described earlier. Results indicate errors of less than 1 part in 109

. The

10:1 and 100:10 ratios were compared to the 10:1 ratio of a Harvey-type comparator [3] accurate to 4 parts in 1010

. Measurements were limited to 5 parts in 109 because

of noise and resistor instabilities. BRIDGE

Fig. 3 shows the bridge connection of the divider resistors (NR and R) and comparator for the purpose of calibrating the divider and measuring the power coefficient of these resistors. A superconducting switch [8] is used

to switch the system from this configuration to a simple

series connection of the resistors as a voltage divider. The

windings of the comparator are superconducting; thus,

CURRENT

SUPPLY

Winding Fig. 3. Circuit for the current comparator bridge. The resistors are trimmed so that the error 5 is small (of order 10-6 or less) and thus

the bridge is not far from balance. A current I3 is supplied to a

trim winding to realize the final balance condition. The coils and

all interconnections are superconducting, insuring that identical voltages appear across the resistors.

the voltage drop across each resistor is the same and the

ratio of power dissipations is the inverse of the resistance

ratio. The resistors are initially trimmed to be close to the desired ratio, and the deviation from the divider

ratio can then be measured by noting the relative current

required in the trim winding to achieve null. Thus, if the

initial balance of the resistance ratio is 1 part in 105

, the trim current relative to the main current need be known

to only 1 part in 104 in order to realize a 0.001 ppm measure of the resistance ratio. This level of accuracy can be

readily obtained with a simple room temperature resist- ance network to divide off part of the main current and

deliver it to the trim coil (see [4]). The fact that the

dissipation level in the two resistors is so much different allows one to measure the power coefficient readily. One

simply measures the bridge output as a function of current and observes that a nonlinearity in this relationship

results from self-heating changes in the smaller resistor. RESISTORS

The two-terminal resistors are composed of a silicon- copper alloy [5] (96-percent Cu-3-percent Si-l-percent

Zn). The wire diameter is 0.25 mm, a factor of 2 larger

than that used in previous resistors [4], [9], [10]. This

factor of 2-in diameter is important since, for the same

value of resistance, it provides an eightfold improvement

in performance (because the surface area for heat dissipa- tion is increased by a factor of 8). The resistor terminations are quite similar to those

described by Harvey and Collins [4]. The wire is hard

soldered (50-percent Ag-50-percent Cu) to a piece of copper, and connection to the superconducting circuit is then made with a lead-tin solder connection to the copper

block. The direct application of solder to the resistance wire results in an added problem which is especially acute

679
SULLIVAN AND DZIUBA: COMPARATOR BR1DUK

for short (low-value) resistors. The boundary between

superconducting and normal metals can move slightly

with changing current and thus causes an effective change

in resistance. Resistors described earlier [9] were ~7 m

long and this problem was not too severe. However, the

resistor described in this paper is ~16 cm long and

with direct solder connections was found to vary at low

currents by more than 1 ppm. The form of this variation involves a rapid change with changing current at low

current levels with a saturation at higher current; a dis- tinctly different behavior than produced by heating. The

use of the copper intermediary terminal seems to eliminate

this problem.

The bridge measurements were performed in a fashion

similar to that described by Harvey and Collins [4]. Referring to Fig. 3, the balance equation is Nh = h + h. Then noting the conservation of current I0 = I\ + /•_> and the equivalence of the voltages across the resistors we have

neglecting the second- and higher order terms of 5. Thus 8, the correction to the resistance ratio, is determined by the comparator ratio JV and the accuracy of the ratio h/hThe accuracy with which 8 is measured is, of course, also dependent upon the sensitivity of the null of the bridge.

The implicit assumption for the approximation which

leads to the last equation is that the value of 8 is smaller than the precision to which it must be measured. Thus, if the resistors are trimmed to roughly 1 part in 105

, then

the last equation can readily be used to determine 8 to 1 part in 104 and a divider calibration of 0.001 ppm results. The resistor power coefficient is then measured by noting

variations of 8 with the current 72 . The bridge output

when plotted as a function of h (which is proportional to h) is a quadratic function of current, consistent with

self-heating. The power coefficient is plotted in Fig. 4 as a

function of temperature. This particular plot is for the

1-U resistor which is expected to be used at a power level of 0.1 mW. Harvey and Collins [4] have obtained a power coefficient measurement for this same material (0.125-mm diameter rather than 0.25-mm diameter) at

2.1 K and it is worth comparing these results with their measurement. At 2.1 K, Harvey and Collins measured

the power coefficient to be —0.013 ppm/mW for a 10-ft resistor. From Fig. 4, the corresponding coefficient is — 0.023 ppm/mW ( 1-12 resistor) which implies a coefficient of —0.0023 ppm/mW for a 10-ft resistor. If the wire diameter were reduced by a factor of 2, the surface area

for heat dissipation would be reduced by a factor of 8 and the power coefficient should be —0.0023 X 8 = — 0.0184 ppm/mW which is reasonably consistent with

the Harvey and Collins measurement of —0.013 ppm/mW.

Some difference might well be expected since the insulation

TEMPERATURE (Kelvin) Fig. 4. Power coefficient of the 1-S2 resistor as a function of temperature. Below the X point (2.17 K), heat dissipation is limited solely by the boundary conductance between the wire and superfluid. Above the X point, strong convection limitations exist producing an increase in power coefficient.

was slightly different in thickness on the two wires and

the extra reduction in diameter might be expected to

affect the temperature coefficient of resistance. While we

have not measured the temperature coefficient of resist- ance of this particular sample of wire, the agreement with

the Harvey and Collins number suggests that the temperature coefficient is probably near that which they

measured, namely, —8 ppm/K at 2.2 K. Another important facet of the operational characteristics of the resistors is the differential temperature

coefficient of two resistors since this determines the bath

stability required for a desired ratio stability. We simply

note here that the resistance ratio varied by less than

0.2 ppm between 4 and 1.5 K while the absolute resist- ance changes by more than 8 ppm. Thus the temperature

coefficients appear to track within a few percent and the bath stability requirements are reduced significantly (a bath stability of 10 mK results in better than 0.001 ppm

ratio stability). These two resistors were fabricated from

adjacent segments of the resistance wire and perhaps

this is necessary to assure a small differential temperature

coefficient. The thermal boundary resistance was calculated [4] to be 0.2 W/cm2 K at 2.1 K, a result which falls within the range of values reported for uncoated copper surfaces

[11]. Thus one gains confidence that the resistor power

coefficient is determined solely by the boundary heat

transfer and the temperature coefficient of resistance. This

confidence is needed to make the assumption that the power coefficient of the wire in the larger resistor ( 100 Q

in this case) is the same as that in the smaller one (1 fl). This assumption is valid only below the X point ( 2.17 K)

since it is only there that the boundary resistance dominates the heat transfer. Indeed, the large increase in power coefficient at the X point is a result of the onset

680
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER l'J74

of convective flow limitations which accompany the disapearance of the superfluid. The boundary conductance

continues to increase (proportional to T3

) with temperature above the X point.

CONCLUSIONS

The development of superconducting current comparators has progressed well beyond the accuracy require- ments for most applications. While these comparators

are orders of magnitude more sensitive than their conventional counterparts, one could argue that further improvement would be of value. Present comparator

sensitivity is limited by noise in the RF preamplifier of the SQUID system and thus further improvement of the comparators should accompany future reductions of this noise contribution. The overwhelming advantage of a comparator-calibrated

resistive divider is the availability of a completely consistent set of internal checks on the system performance.

The comparator windings can be intercompared to as- certain the ratio uncertainty, and the power coefficient of the resistors can be determined directly during the

divider calibration. The operation of a resistive divider above the X point

offers a simplification which might be attractive. The

value of the power coefficient at 4 K given by Fig. 4

results in an error of 0.01 ppm at 0.1-mW dissipation, a

not unreasonable number. Unfortunately, one cannot assume that the power coefficient for the 100-12 resistor

is the same. In fact, it will likely be larger since the wire

is more closely packed as a result of its length, and convection of heat by the fluid is thus poorer. This matter

could be checked by using a still larger resistor as a refer- ence in measuring the power coefficient of the 100-12

resistor. In any event, further work is necessary before such resistors can be used above the X point. The development of a resistance material with a much smaller temperature coefficient, while not essential, would certainly

offer welcome simplifications.

REFERENCES

[1] I. K. Harvey, "A precise cryogenic dc ratio transformer,"

Rev. Sri. Instr., vol. 43, pp. 1626-1629, Nov. 1972.

[2] K. Grohmann, H. D. Hahlbohm, H. Gubbig, and H. Ramin,

"Construction principles and properties of ironless dc and ac current comparators with superconducting shields," presented

at the Electrical and Electronic Test Instrument Conf., Ottawa, Canada, May 1973; also, in PTB-Mitteilungen, vol. 5, pp. 313-

318, May 1973.

[3] D. B. Sullivan and R. F. Dziuba, "Low temperature direct current comparators," Rev. Sri. Instr., vol. 45, pp. 517-519,

Apr. 1974.

[4] I. K. Harvey and H. C. Collins, "Precise resistance ratio measurements using a superconducting dc ratio transformer,"

Rev. Sri. Instr., vol. 44, pp. 1700-1702, Dec. 1973.

[5] D. B. Sullivan, "Resistance of a silicon bronze at low temperatures," Rev. Sri. Instr., vol. 42, pp. 612-613, May 1971.

[6] J. E. Zimmerman, "Sensitivity enhancement of superconducting

quantum interference devices through the use of fractional-turn loops," J. Appl. Phys., vol. 42, pp. 4483-4487, Oct. 1971.

[7] J. E. Zimmerman, P. Thiene, and J. T. Harding, "Design and

operation of stable rf-biased superconducting point-contact

quantum devices, and a note on the properties of perfectly clean metal contacts," J. Appl. Phys., vol. 41, pp. 1572-1580, Mar. 1970.

[8] J. D. Siegwarth and D. B. Sullivan, "A mechanical supercon- ducting switch for low temperature instrumentation," Rev. Sri.

Instr., vol. 43, pp. 153-154, Jan. 1972.

[9] D. B. Sullivan, "Low temperature voltage divider and null detector," Rev. Sri. Instr., vol. 43, pp. 499-505, Mar. 1972.

[10] J. C. Gallop and B. W. Petley, "Recent NPL work on the AC

Josephson effects as a voltage standard," IEEE Trans. Instrum. Meas., vol. IM-21, pp. 310-314, Nov. 1972.

[11] N. S. Snyder, "Thermal conductance at the interface of a solid and helium II (Kapitza conductance)," Nat. Bur. Stand., Tech. Note 385, Dec. 1961.

Reprinted by permission from IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT

Vol. IM-23, No. 4, December 1974, pp. 256-260 Copyright 1975 by the Institute of Electrical and Electronics Engineers, Inc. PRINTED IN THE U.S.A.

681
JOURNAL OF RESEARCH of the Notional Bureau of Standards — C. Engineering and Instrumentation

Vol. 75C, Nos. 3 and 4, July-December 1971 Thermal Voltage Converters and Comparator for Very

Accurate AC Voltage Measurements

E. S. Williams

Institute for Basic Standards, National Bureau of Standards, Washington, D.C. 20234

(July 22, 1971)

A new fourteen-range set of thermal voltage converters and a thermoelement comparator are used

to measure ac-dc difference, and a-c voltages relative to external d-c standards, with 20 ppm (parts- per-million) accuracy at audio frequencies. The imprecision is less than 2 ppm. Corrections relative to the very stable middle ranges can be redetermined for every range by a seven-step intercomparison of certain adjacent ranges.

Key words: AC-DC difference: comparator: thermoelement: transfer voltmeter: voltage measurements.

1. Introduction

An a-c voltage at audio frequencies is measured

most accurately at the present time by using a thermal

voltage converter (TVC) to compare it with a stable and accurately measured d-c voltage, which is nominally equal to it. The basic d-c standards are then,

in effect, "transferred" to the a-c measurement. The

TVC may be simpiv a thermoelement (TE) in series with an appropriate multiplier resistor. The output

emf of the TE is ordinarily monitored with a nuD

detector and a balancing circuit, which may be a Lindeck potentiometer. A balancing circuit and a null detector are included in most commercial multi- range models.

Before the TVC is used for a-c voltage measurements

it must be tested for ac-dc difference or frequency in- fluence so that corrections may be applied. The set of TVCs and the new TE comparator described in this paper were developed primarily for making these

tests. However they may also be used for a-c voltage

measurements as explained in section 9. For ac-dc

difference measurements the TE comparator, whose

read out instrument may be either a nanovoltmeter or a galvanometer, provides a considerable advantage over

other methods in overcoming the difficulties caused

by power supply instability and inexact voltage control. The 14-range set of TVCs (designated No. 7) consists

of six resistor units and two TEs, and extends from 1 to 1000 V. The TEs are rated at 2.5 and 5.0 mA, and each one may be attached to any one of the resistors by a coaxial connector (see fig. 1). This permits each resistor

to be used for two voltage ranges, as shown in table

1. Certain adjacent ranges of the set can be inter- compared, and the ac-dc differences of all the ranges

can be determined relative to any one range. This

set is otherwise similar to an earlier set [1J

1

, which

consisted of two 5-mA TEs and 12 series resistors,

and covered a range from 0.5 to 500 V. The earlier set, designated No. 1, has been extended to 1000 V with one additional resistor as part of the present project.

It is evaluated in a 14-step intercomparison in which

each range is compared, at reduced voltage, with the next lower one.

In a well designed TVC the ac-dc difference is not affected by changes in the applied voltage. Therefore, an ac-dc difference determined for the 100-V range at 60 V can be applied as a correction when this resistor

is used at 200 V to test the nominal 300-V range of the

next higher resistor. Experience has shown this stepup procedure to be feasible. However if the higher volt- age ranges are not carefully built the ac-dc differences can change due to self-heating effects to be described

later.

2. Design and Construction

The two TEs are mounted in 2-in brass tubes

(see figs. 1 and 2, upper right) with coaxial connectors

for attaching to a resistor. The 2.5 mA-TE has a 400 O

heater and is used alone as the 1-V range. The 5 mA-TE

has a 125 ft heater and a 275 ft resistor is added inside

its enclosure to make a 400 Q, 2-V range. The middle ranges (six ranges, 3 to 60 V) make use

of three resistors — 0.8 kH, 3.6 kil, and 12 kii (see table 1). Each of these is a 2W metal-film resistor mounted coaxially in a 3-in brass cylinder 4 1/2 in long with coaxial connectors at each end.

Figures in brackets indicate the literature references on page 153.

682
FIGURE 1. New set of thermal voltage converters. The simple symmetrical geometry of these units

permits approximate calculations of their reactances

[2]. The calculations are inexact because of the nec- essary assumptions as to end-effects, and they neglect

any small residual reactances of the resistors themselves. They do indicate however, that the frequency

error of each resistor unit (without the TE) should be

less than 1 ppm at 50 kHz, even for tubes smaller than

2 in in diameter. (The 3-in tube was chosen for the

higher ranges where more space is necessary and was

therefore used for the middle ranges also.) The higher voltage ranges have larger resistor as- semblies to avoid excessive temperature rise. The

100-200 V unit has a 40 kfl spiraled metal-film resistor

five inches long. 2 The 300-600 and 500-1000 V units

have resistors of tin oxide deposited on a 0.81-in glass

tube 6 in long. A small voltage coefficient was antici- pated in these resistors, but measurements have shown

it to be negligible.

This experimental resistor is not ordinarily available commercially. Four 40 kO metal

film resistors in series-parallel can also be satisfactory.

TABLE 1. NBS thermal voltage converter set No. 7

Fourteen voltage ranges (column 2) are formed with six series

resistors and two TEs iF 1 and F 2).

Series Voltage TE Total

resistor range resistance mA m

1 2.5 (F 1) 0.4 2 5.0(F2) 0.4

3 2.5 1.2 0.8 6 5.0 1.2 10 2.5 4.0 3.6 20 5.0 4.0 30 2.5 12.4 12 60 5.0 12.4 100 2.5 40.4 40 200 5.0 40.4 300 2.5 120.4 120 600 5.0 120.4 500 2.5 200.4 200 1000 5.0 200.4

Errors in these ranges are caused mainly by capacitance between the resistor assembly and the outer

casing, which permits alternating current to bypass

the TE to ground. Therefore more a-c than d-c voltage

is required for a given TE output. Frequency compensation could be provided by placing relatively small

capacitors in parallel with part of the resistance.

However, such capacitors probably would not be suf- ficiently stable over a long period of time and might

be affected by temperature changes which occur in the

resistor enclosure.

Compensation was therefore provided (as in the

earlier set No. 1) with an inner shield which is con- nected to the input and surrounds the high- or inputend of the resistor (see "S" in fig. 2). The shield is positioned, relative to the resistor, to control the capacitance currents and provide optimum high-frequency

FIGURE 2. Essentials of thermal voltage converters.

Thermoelement (TE) enclosure, low voltage unit with resistor |R), and high voltage

unit 1 100 to 1000 V) with resistor iR) and frequency compensation shield (S).

683
compensation: perfect compensation is possible only

at one frequency.

3. 1000-VTVC

As stated earlier the corrections for the high and low

voltage ranges are determined relative to the middle

range by intercomparison tests. It is necessary there- fore that the ac-dc difference of all ranges be unaf- fected by voltage level. The voltage effect is rare in low

and middle ranges, but it can be troublesome at higher

ranges where the heat generated by the resistor is appreciable. All intercomparison tests were initially

made at two voltage levels in order to evaluate this effect — or to prove its absence. If, for example, the

nominal 600-V range is compared with the 1000-V

range at both 400 and 600 V, and the same results are obtained, it is unlikely that the 600-V range is affected

by the voltage increase (the 1000-V range is very

probably unaffected at these voltages). However both

ranges are in question until we are assured that the

1000-V range is also free of this effect. The highest range, then presents a special problem.

It is more likely to change due to increased voltage

(heating) and there is no higher range to compare it with. If it is compared with another 1000-V TVC at two or more voltages and no change is observed it is possible that both have the same voltage effect. However this is unlikely if TVCs are of somewhat

different design. If intercomparisons of several units of difference design show no change between rated

voltage and a lower voltage we may safely assume

this effect is negligible in each one. Five 1000-V TVCs (table 2) have therefore been

built, each with differing components, and a network

of intercomparison tests was made as described in the next section. The original set of TVCs [1] was included

in these tests. Table 2. Five 1000-V TVCs

Unit Resistance Voltage range with TE Resistor assembly

F 1 F2

A

Afl 400 1000 Four 100 Afi wire- wound resistors in series, each 1 3/8 in long.

B 400 1000 Four 100 AH wire- wound resistors in series, each 1 3/8 in long. C 200 500 1000 Four 200 AO metalfilm resistors in series-parallel, each

2 in long. D 200 500 1000 One 200 Afl tin oxide

resistor 6 in long.

E 400 1000 Four 400 Afl metalfilm resistors in series-parallel, each

2 in long.

The five 1000-V TVCs are designated alphabetically

(A through E)3 and all except "A" have the inner shield, as described below, for frequency compensation. The

resistor in unit "A" is mounted concentrically in a set of six brass rings whose potentials are maintained by a capacitance divider. The whole assembly is mounted in a 4-in brass cylinder 7 in long.

Self-heating of a TVC may affect either the fre- quency compensation or the resistor itself. Apparently

the dielectric losses in the insulation between con- ductors can be affected by heating and change the

impedance of the resistor. This change occurred within

a few minutes after a voltage increase or decrease in several resistor types, which were therefore discarded.

The frequency compensating shield may be moved,

relative to the resistor, by thermal expansion of mounting parts and cause a change in ac-dc difference. This

occurred in some of our early units and the change was slow, taking up to an hour. Later shields were

therefore mounted very rigidly since a small displacement will have a large effect on the frequency influence.

The shield now in use is in two parts. A brass

cylinder, with one end closed, is mounted firmly

against the outer end piece (input end) with a polystyrene insulator one-half inch thick (fig. 3). A movable

tube fits tightly inside this cylinder and the shield

length adjustment is made by pushing this piece for- ward with a small rod inserted through the end piece

and the insulator. The rod is removed after each

adjustment.

The shield is most effective at the leading edge

where the voltage difference is greatest between the

resistor and the shield. Therefore the movable part of the shield is cut at an angle, so that only part of it (see fig. 3) extends outside the fixed cylinder. Thereby

the effect of small position changes is reduced. Even

so, the compensation is affected by approximately 0.01 percent per millimeter of position change in some

typical units. The shield length adjustment is made in small steps,

and tests are made after each adjustment. When

optimum compensation is achieved, the unit is opened

and the shield parts are locked together with three screws through their overlapping portions.

4. TE and TVC Tests

As stated earlier the middle ranges are the most

accurate, and they are the base from which higher

and lower ranges are tested. However they are no more accurate than the TE used with them. Before

being installed in a TVC the TEs are therefore compared with one of a group of carefully selected TE stand- ards which were made according to NBS specifications.

This group contains twelve elements made by four

manufacturers, and are rated from 5 to 20 mA. [1] They

have either Karma or Evanohm 4 heaters to reduce

3 Unit "D" is assigned to this set (No. 7). •Certain commercial products and instruments are identified in this paper in order to specify adequately the experimental procedure. In no case does such identification imply recommendation or endorsement by the National Bureau of Standards, nor does it imply

that the products or equipment identified are necessarily the best available for the purpose.

684
Figure 3. Cross section of high voltage TVC.

Input end is at the left. Fixed and adjustable shield parts are supported by a polystyrene insulator one-half inch

thick.

thermoelectric effects in the heaters. They also have

small reverse d-c differences (less than 200 ppm).

high bead resistances (over 1000 Mfi). and small ac-dc

differences. An additional two are multi-junction

elements of different design. All of these TEs have

been intercompared as current converters with agree- ment to better than 3 ppm. Theoretical calculations,

taking into consideration all known sources of error, also indicate that the error should not exceed a few

ppm at audio frequencies. Because of this and because

of the excellent agreement between these thermoelements of different design and construction, we may safely assume that the average ac-dc difference

of the group is 0 to ± 2 ppm.

The low-frequency performance of a TVC depends

on the TE since the reactance of the resistor is entirely

negligible even at a few kilohertz. The accuracy of

a TE at low frequency is mainly dependent on the

length of the heater. A very short heater permits heat

to flow more readily to the support stems and cooling

to occur between current peaks. However most commercial TEs have errors less than 10 ppm at 20 Hz

and some have errors less than 10 ppm even at 5 Hz.

In testing selected commercial TEs for the TVCs

at audio frequencies we often find a small ac-dc

difference (up to 4 ppm) which is independent of frequency but dependent on heater current. It or- dinarily decreases at lower heater currents. When

these TEs are used in a TVC the ac-dc difference

will therefore be slightly voltage dependent. In the step-up test process, where the TVC is alternately

used at 50 and 100 percent of rated current, this rela- tively small thermoelement error is introduced at every step of the process. If the process is a long one,

as in the 12-resistor set, there could be a significant

accumulation of error. TVCs in the new set are inter- compared by testing each resistor once with one TE

at reduced current (60 or 67%), and again with rated current on the other TE. If these TEs differ in ac-dc

difference at these currents, there will again be an accumulation of error. This is one of the advantages of the set with only

six resistors. If the 20-V range is the starting point

(it is probably the most stable), we have only four

steps up to the 1000-V range, and three steps down

to the 1-V range. The other advantages are that less time is required for the intercomparisons and there

are fewer units to construct.

Intercomparison data for this set of TVCs are listed

in table 3 as well as determinations made using TVC

set No. 1. (Test methods are discussed in later sections

of the paper). The middle ranges differ by 2 ppm or less even though the resistance varies from 1200 to 12000 fl. Since the resistance enters into the theoretical calculation of ac-dc difference, this agreement is an additional indication of accuracy. Step-down tests show that the two lowest voltage

ranges usually have a positive correction (i.e.. more

a-c than d-c voltage is required for a given TE output)

of 5 ppm or less at 50 kHz. There is no resistor in series with the TE at the 1 V range, and at the 2-V

range the TE is a large part of the total impedance.

These ranges are therefore affected more by the small reactance of the TE. The corrections for these ranges

are more likely to change if a TE is replaced than for the higher ranges. A step-down test from the middle

range is therefore advisable when TE replacements

are made.

The five 1000-V TVCs were compared with the

300- and 500-V ranges of TVC set No. 1 and the 600-V

range of the new set No. 7. The corrections are listed

in table 4 for 20 and 50 kHz and all values are averaged.

Table 3. AC-DC difference AC-DC difference (ppm 1 Applied TVC-a TVC-b voltage 20 kHz 50 kH2

8,-8,, 8, 8 8„-8/, 8» 8

Fl 1 -3 -3 + 1 -3

F2 Fl 1 0 F2 2 — 2 -2 4-1 —2

Fl-3 F2 2 - 1 Fl-3 3 - 1 + 1 0 —2

F2-6 6 -1 0 0 -3

Fl-10 F2-6 6 + 1 + 1 Fl-10 10 0 + 1 + 1 + 1 F2-20 20 0 0 + 1 + 1 Fl-30 F2-20 20 -2

Fl-30 30 - 1 + 1 -1 +2

F2-60 60 - 1 0 -1 0 Fl-100 F2-60 60 -4

Fl-100 100 -2 0 —5 —2

F2-200 200 — 2 -1 —5 -6

F 1-300 F2-200 200 + 6 F 1-300 300 -4 -3 + 1 -4

F2-600 600 -4 -4 + 1 -1

Fl-500 F2-600 500 -26

Fl-500 500 -6 -3 4-27 -30

F2-1000 1000 -6 +27

685
Table 5 lists intercomparison data for the same fre- quencies for seven pairings of the five 1000-V TVCs.

Values for 600 and 1000 V do not differ significantly and

so they are all averaged together.

Corrections (5) for each range of the new TVC set are listed as determined in comparison with TVC Set No. 1. Intercomparison data(8„ — 8b) are also listed and

a computed correction (5S ) based on the intercomparison. starting with the average of the measured ac-dc

differences of the 10- and 20-V ranges. The TVCs are

identified as the combination of a TE (Fl or F2) with

the appropriate resistor: i.e.. F2-600 forms the 600-V

range. Averages of the corrections for the five 1000-V

TVCs listed in table 4 are shown in the circles in figure

4, and the connecting lines show the intercomparison

values from table 5. The agreement between the vari- ous units is illustrated by this figure.

Each of the values listed in tables 4 and 5 is the

average of at least four separate determinations.

An analysis of the data taken in a series of 26 inter- comparison tests (an average of 5 determinations

with each one), involving all ranges at two frequencies and usually two voltages, showed an imprecision

(3 o-„, where aa is the standard deviation of the aver- age of five determinations) of only 1.4 ppm.

Table 4. Ac-dc difference corrections for five 1000-volt TVC (A-E)

The standard instruments are the 300- and 500-V units of Set No. 1 and the 600-V range of Set No. 7.

Standard Volts A B C D E AC-DC difference at 20 kHz (ppm)

5O0-1 500 -20 -9 -4 -12

300-1 300 -4 -10

600-7 600 -3 -22 -2 -12.-10

-8 -3 -10

Ave. -6 -21 -9 -3 -11

AC-DC difference at 50 kHz (ppm)

500-1 500 + 21 -6 + 11 + 30 -3

300-1 300 + 30 -4

600-7 600 + 30 -9 + 30 -6

+ 23 + 32 -6

Ave. + 25 -7 + 11 + 30 -5

Table 5. Relative ac-dc differences for seven pairings of members of

the five 1000-V TVCs

Volts sA_sB 8B-8C 5C-5D 8d~Oe 8A -5C 8a_

8d AC-DC difference at 20 kHz (ppm)

600 -11 + 14 + 2 -7 + 7 + 3 0

1000 -17 + 13 0 -6 0 + 9 + 5 600 -16 + 17 -10 -2

1000 -17 + 5 Ave. -15 + 15 + 1 -7 + 4 + 6 + 2 AC-DC difference at 50 kHz (ppm)

600 -20 + 14 + 10 -33 + 28 -15 + 3

1000 -22 + 6 + 2 -33 + 35 -14 + 4 600 -25 -37 -15 + 2 1000 -24 + 1 Ave. -23 + 10 + 6 -34 + 32 -15 + 2

5. Thermal Compensation

The coaxial connector between the TE and the series

resistor provides a low-reactance connection for minimum frequency error. Tests and calculations show the

error is less than 1 ppm at 50 kHz. However, it also permits heat to flow freely from the resistor, which

dissipates up to five watts on the 1000-V range, to the

TE which has a temperature coefficient of emf of ap- proximately 0.2 percent/0 C (at constant input current).

Heat conduction through the center conductor of

this connector was reduced, at 300 V and higher, by

replacing the two polystyrene insulators in the coaxial connectors (one in the resistor output and one

in the TE input) with two made of boron nitride. This

material is a good electrical insulator and yet it con- ducts heat well enough to provide an effective heat

sink for this conductor. It is used in both ends of the

high voltage resistors to improve the heat flow from the

resistor to the relatively heavy brass casing. This also helps to reduce the temperature rise in the resistor. Heat flow through the outer part of these connectors

will gradually raise the temperature of the TE casing,

and this also causes a heating drift. The increase in temperature of the TE bulb is slowed by mounting it in a short section of brass tubing to increase the

thermal mass and this tube is thermally insulated from

the TE casing. Thermal compensation is also added by

attaching a 10-O thermistor to the tube containing

the TE and connecting it in series with the TE output.

A resistor appropriately chosen to complete the compensation circuit (see Fig. 7 and appendix I) will draw current through the thermistor sufficient to compensate for the temperature increase.

This resistor value may be computed for TEs with

average temperature coefficients as in appendix I. However, temperature coefficients differ considerably

and. for best results, it is usually necessary to match

the resistor to the TE and thermistor combination. A

suitable value can be found by substituting a variable

resistor outside the TE casing and adjusting it for minimum warmup drift on a 1000-V range. However, if the

TVC is to be used with a TE comparator similar to the one to be described, it will frequently be connected

20 kHz 50 kHz

FIGURE 4. Intercomparisons of 1000-volt TVCs.

Ac-dc difference corrections Ippml from table 4 for five 1000-V TVCs are shown in the circles and the relative differences from table 5 are shown on the interconnecting

lines. The algebraic difference between anv pair should be equal to the figure on the connecting line within experimental erTor.

686
across a 1000-fl divider, and this will affect the compensation by about 20 percent. A value about 10 per- cent higher than the one chosen experimentally should

provide a satisfactory compromise and reduce the

drift by a factor of five or more. With the compensation

and the large thermal time constant of the TE in iis casing, the drift can be reduced to less than 5 ppm/

minute under typical laboratory condition. The error from this is almost completely eliminated by taking a sequence of readings with a-c. d-c, reversed d-c. and a-c voltage applied, as described in the next section.

6. TVC Test Methods

A brief review of other comparison methods may be

useful before describing the TE comparator. Figure 5 shows two TVCs connected in parallel. The separate

balancing circuits (or Lindeck Potentiometers) (B) and

null detectors (D) may or may not be built in. Stable a-c and d-c voltage supplies are connected to the two TVCs

by the ac-dc switch. The switching arrangement shown

is typical and convenient, although others, such as re- versing the d-c supply by a separate switch, are satis- factory. The d-c supply must have an ungrounded

output and it should be possible to ground either ter- minal as the voltage is reversed. The reversed d-c dif- ferences of the TVC can rarely, if ever, be neglected

in these tests. For convenience and clarity, we shall call one TVC

the "standard" and the other the "test." Normally the one with known ac-dc difference corrections is the

standard, and the test TVC is usually treated differ- ently in the test procedures, as explained below.

There are two procedures for this intercomparison.

One might be called a null-balancing method. With a-c voltage applied to both TVCs. the balancing circuits of the test and standard instruments are adjusted to null the detectors. The d-c supply is then switched to the

TVCs. and the voltage is adjusted to obtain a null bal- ance first on the detector of the test instrument and

then on the standard. Each d-c voltage is carefully

measured, and the small difference between the two

should equal the difference between the a-c voltage and

one direction of d-c voltage. The measurement is then

repeated with reversed d-c voltage, and an average is taken.

This method is tedious and time consuming, an ac- curate d-c voltage standard is required, and if there is any drift in either instrument output due to warmup or other reasons, as there very often is. the data may lack

both accuracy and precision.

A more satisfactory procedure and one that will minimize the effect of drift may be called a deflection

method. Both balancing circuits are adjusted to near

null, and then a-c and d-c voltages are applied in the

succession indicated in figure 5 (AC. DC+. DC—, AC).

For each input the power supply is adjusted to set the

test instrument detector to null. The deflection of the

standard TVC detector is read for each setting, and the

difference between the average reading on a-c voltage

and d-c voltage is an indication of relative ac-dc differ- ence. If the readings are made at nearly equal time

intervals, and if the drift rate is reasonably constant

(even though fairly large), the determinations can be

made with satisfactory precision.

If the detector of the standard TVC is a galvanometer

the reading may be in divisions or millimeters, but they can be converted to volts or percent ac-dc differences

by calibrating the detector scale in one of several ways.

A calibration factor can be determined simply and di- rectly by making a small measured change in the d-c input and observing the detector response.

If the detector is a microvoltmeter and if the balancing circuit is a Lindeck potentiometer indicating the

millivolt output of the thermoelement, the calibration

step can be eliminated. With the four deflection readings on the detector the difference between the average

reading on a-c voltage (E„) and those on d-c voltage

(E,i) is computed from the differences of the corresponding readings of the microvoltmeter. The ac-dc differ- ence of the test instrument is

S

t

. Esa Esd

t — Os H -p,

where the subscript s indicates that these values come

from the standard instrument, and 8S is the ac-dc differ- ence correction for the standard.5 The factor n relates small changes in thermoelement

heater current (A/) (at a fixed frequency) to corresponding changes in output emf (A£/£'= nA/// approximately). For a TVC we may substitute nAV/V for the right

hand expression, so that n= VAE/EAV. The value of n is 2 if the thermoelement has a squaie law response,

possibly at very low heater currents, but it is usually

1.7 to 1.9 at rated current. Determinations of n should

therefore be made in a d-c test at five or more current

levels by measuring the values indicated. A plot of n

against E can be made so that values corresponding to any emf can be found for substitution in the equation

above (see sec. 8).

FIGURE 5. Two-P'otentiometer method for TVC comparisons. 5 See eq 3 of appendix II. This general formula is applicable here also.

687
7. TE Comparator

There are several variations of basic comparator

circuits which are used to minimize the difficulty

caused by power supply instability [1. 3]. They employ

a voltage divider circuit to which two emfs are con- nected. The divider is adjusted to null a detector,

and at this point the divider setting corresponds to the ratio of the emfs. If one emf (E,) is held constant

as a-c and then d-c voltage is applied to the TVCs,

the other emf (E«) will change if there is a relative ac-dc difference in the TVCs at the frequency (8, 8S ) The resulting unbalance in the divider will produce a change in the detector deflection proportional to the

ac-dc difference. However, small fluctuations in the

power supply will produce nearly equal proportional

changes in the emfs and therefore the detector will not be affected appreciably. The stabilizing effect depends on how well matched the time constants and response characteristics of the TEs are. but the

effect usually affords a significant advantage. Also,

the monitored emf need not be held constant so exactly as with a simple balancing circuit. In the new comparator (fig. 6) the higher of the

two emfs is connected across a 1000-fl Kelvin-Varley

divider, and the lower one is connected to the variable

tap. The detector is brought to null by adjusting the

divider. A Lindeck potentiometer and the same de- tector (with key ki) is used to monitor the test TVC

output (Ei). The detector is labeled "N" in the dia- gram because the use of a nanovoltmeter is suggested.

The test procedure is similar to that already mentioned. Preliminary settings of the potentiometer

and divider are first made so the readings of the

nanovoltmeter will fall near mid-scale. Then with a-c voltage applied, the key ki (labeled "SET" on the

instrument, figs. 8 and 9) is depressed and the approI

FIGURE 6. New TE comparator.

priate power supply is adjusted for a null on the

detector. Key kj (READ) is then depressed and the

detector indication is read. The two-step procedure

is repeated for two directions of d-c voltage and for a second a-c input. The average indication in nanovolts K (N„) for a-c input and the average for d-c input

(/Vrf) is computed and the ac-dc difference of the

test instrument is

5r = 8v+ ^d J^" (appendix II eq 5)

Figure 6 shows the test TVC output (Et ) connected

to the high emf terminal (Eh in fig. 8). If E, is smaller

than E* it will be necessary to interchange them, but

this is a minor inconvenience. If E, is connected to the E[ terminal the potentiometer is switched to that input with switch Sj (fig. 8) and E, is monitored as

before. The same switch also reverses the polarity of the detector, and therefore the sign of the indication. The slightly modified equation for 8( is eq 11 in appendix II. A galvanometer may also be used as a detector but

since the circuit resistance is relatively high a photoelectric amplifier is necessary for sufficient sensitivity.

The scale must be linear over the portion to be used

and a scale calibration is also necessary to relate the

readings in divisions or millimeters to a voltage change.

The calibration can be made quite easily however by

moving switch Si from READ to CAL with k2 closed

and observing the resulting deflection change. This

increases the resistance of the lOOO-H divider by 0.5 Cl and changes the divider current by 0.05 percent. The

SET SEAO

Figure 8. New TE comparator

B 6< — ft, is convenienlly in parts-per-miiJion it E. is in millivolts and Nh — .V., is in nanovoits.

688
FIGURE 9. Photograph of TE comparator.

voltage applied to the galvanometer circuit is changed

by the same amount and the deflection change D\

is used in eq 10 and 12 in appendix 2 to determine

8f . A measurement of Es is made only to find the cor- responding value of ns from a plot as before.

8. n Tests

Measurements of n are conveniently made with the

comparator. The thermoelement to be tested is con- nected to the EL input and the EH input is unused. A

key labeled "n TEST" (fig. 6) is locked down and

switch S2 is moved to EH connecting the potentiometer

voltage across the divider. The divider is set arbitrarily but preferably near full range and the potentiometer is adjusted to null the detector. The input

voltage is then changed by a small measured amount

&V/V, and the resulting change in TE output (A£)

is measured by observing the change in indication of

the detector (nanovolts or microvolts). 7 Switch S-> is then moved to EL and a measurement of TE output

(E) is made with the potentiometer. The values for each factor of the equation given above (n = VAEI

E&.V) are then known and n is calculated from this equation.

If a galvanometer is used as the detector a measure- ment of Di is made as described above, and the deflection change resulting from the measured change in input voltage {bV) is designated D2. Then rc = 0.05

DzlqD\, where q is the percentage change in input

voltage (100 &VIV).

Determinations of n need be accurate only to a few

percent, and, since they are quite stable, retesting is not usually necessary.

9. AC Voltage Measurements

As suggested earlier, the comparator may be used

with one TVC to measure an a-c voltage such as the

7 A small change in the input to a TVC can be measured with a volt box and potenti- ometer or by a setting with a calibrated d-c voltage supply. It is also convenient to use a

resistor with a shorting switch in series with a TVC to introduce a small current change

(AY 1 in the thermoelement healer (i.e.. 20 n with a 40-kn. 20O-V TVC will give 100 A/// =

0.05 percent which is equivalent to 100 ClVIV) .

input to a digital voltmeter or the output of a calibrated

a-c voltage supply. Switch Si has a third position

marked "POT'R" which disconnects the divider from

the EH input. If a TVC output is connected to EH , and

switch S2 is moved to that input, the potentiometer and

detector can be used to measure the TVC output and

changes in it. The circuit is as figure 5 without the test TVC. The d-c voltage is accurately measured and nominally equal to the a-c voltage.

It is suggested that d-c voltage be applied to the TVC

first and a detector reading taken. Then, at nearly equal

time intervals, a second reading is taken with a-c volt- age and a third with reversed d-c voltage. The differ- ence between the average of the readings with a d-c voltage (Ed) and the one with a-c voltage (Ea ) is related

to the input voltage difference very much as before.

•E, nE

With a nanovoltmeter the emfs are read directly, but

if the detector is a galvanometer a scale sensitivity factor must be determined.

10. Design Details

The case of the TE comparator has thermal insulation and electrostatic shielding, and the comparator is carefully built to minimize thermal emfs and contact

resistances. The switch decks of the Kelvin-Varley

divider and Si and S2 have enclosed silver contacts and

the keys ki and k> are of a low-thermal type. The back

contacts on these keys connect a 100-fi shunt across

the detector to reduce sensitivity and noise until one

key or the other is depressed. The fine control for the

divider is a ten-turn helical resistor with a special lowthermal sliding contact. The resolution of the uncali- brated divider is not quite sufficient for setting an exact

null on the detector, but setting within a few divisions

is adequate in the ordinary use of the instrument.

689
1 1 . Conclusion

The use of one resistor unit for two TVC voltage

ranges has reduced by about half the time required, over previous TVC sets, to make a complete intercomparison test. Changes in any member of the set are

readily detected since frequent intercomparison tests are more feasible. The TE comparator combines the stabilizing fea- ture of other comparators with much of the simplicity

of the single Lindeck potentiometer or balancing

circuit. It is relatively inexpensive to build and calculations are simplified and direct, especially where the

detector is a nanovoltmeter. Repeated tests have shown

that the imprecision is less than 2 ppm and ac-dc transfer accuracies, with the TVC set, are 10 ppm at audio

frequencies and 20 ppm up to 50 kHz.

change in thermistor resistance. Ai? ( = a/? ( /100

where a is the temperature coefficient of the thermistor

in percent/ 0 C — typically about — 4 percent/ 0 C. Therefore, for thermal compensation,

/3 = 100A/?,/7? = aRJR

and R = aRjp

With the typical values listed we find R = 200 Q,

and. subtracting Rc and Rt , the value for Rs is 180 fi.

14. Appendix II

Development of Equations"

The ac-dc difference of a TVC is defined as

This work was supported in part by the Army Metrology and Calibration Center, Redstone, Alabama.

Their encouragement and support are gratefully

acknowledged. The author also acknowledges the

help of C. B. Childers and A. G. Perrey who made

many of the exacting measurements described here.

12. References

[1] Hermach, F. L. and Williams, E. S., Thermal converters for audiofrequency voltage measurements of high accuracy, IEEE

Transactions on Instrumentation and Measurement. IM-15,

No. 4, 260-268, (Dec. 1966).

[2) Hermach, F. L. and Williams, E. S.. Thermal voltage converters

for accurate voltage measurements to 30 megacycles per second.

Trans. AIEE (Communication and Electronics). 79, Pt. I, 200-206, (July 1960).

[3] Turgel, Raymond S., A comparator for thermal ac-dc transfer standards, ISA Transactions, 6, No. 4, 286-292. (1967).

13. Appendix I

The temperature coefficient of emf, /3, of the

average TE is about —0.2 percent/ 0 C, with constant

input current. Thus a corresponding decrease in the resistance, R = Rc + R> + Rs (see fig. 7) is required

to maintain a constant voltage across the resistor Rs . For a 1 degree increase in temperature, the

FIGURE 7. Thermal compensation for TE.

R, TE output resistance (about 1(H1). H, Thermistor resistance labout 10O). R, Series resistor to complete compensation circuit. iRr and /?, have negligible temperature coefficients./

8 = (i)

where Va is the a-c voltage and Va the average of the two directions of dc-voltage required to produce the same output emf. In the circuit of figure 6 the a-c and d-c voltages are adjusted to give the same emf

Et , of the TVC under test, as indicated by a null on

the detector N with ki closed. The same voltages are applied to the standard TVC. Then,

(2)

where V'd

is the d-c voltage required to produce the same output emf of the standard as Va , and the subscripts, t and 5, refer to the test and standard instru- ments respectively. If V'd —Vd <<l, then, closely

enough

0( — os — —77 . V

?rom the definition of n given in the paper

Esa Esd

8i — 8S —

nsS-CE.Sit

With the polarities as shown in figure 6, N = mE, — Es

(3)

(4)

where N is the detector voltage with k> closed and

ki open, and m is the divider ratio (with the detector

resistance Rd >> R\ )

Thus.

8,= 8S +

Nd -Nn nsEsd

where the subscript a and d have the same meaning

as before.

These equations were developed bv F. L. Hermach.

690
If a galvanometer is used as the detector, the current through it with lt2 closed is, by Thevenin's Theorem,

Igb —

rriEt — Es Re

(6)

where Re is the resistance of the galvanometer circuit with E, and Es = 0. With E, and Ex held constant and

pRi inserted by means of Si,

Igc mE,l(l + p)-Es r:

(7)

where R

'e is the resistance of the galvanometer circuit. Since mE, — Es «Es and p << 1, we have to a

sufficient degree of approximation,

j —i — H — igb ' ac r Esp _ (Db — Dc Re s

(8)

where Dt> and Dc are the resulting galvanometer deflections and S is the galvanometer current sensitivity. In the ac-dc test, with mEt constant,

Iga Igd —

Esd Esa Da ^d

Re S

Thus, from (3), (8) and (9), p(Dd-Da)

8,= 6s +

; nsDi

(9)

(10)

The test and standard TVCs may be interchanged

if E, < Et. If Et is applied to the EL input of figure 8 and

held constant, with the detector and potentiometer con- nected as shown in the figure, a similar analysis leads

to the following equations:

8( = 5S +

Nd ~Nq

risEi

and

8, = 6,+ p{Dd -Da ) nsD,

The characteristic n is determined from

n =

AEIE

LVIV

(11)

(12)

(13)

by applying known changes in input voltage and

observing the changes in output emf with a high resist- ance voltmeter, as described in the text. If a galvanometer is used as the detector instead of the voltmeter

its sensitivity is determined by inserting pR\ as de- scribed. From equations similar to (8) and (9)

&Es _pM)

Es ~ Dx (14)

If the thermocouple resistance of the test TVC,

Ric, is significant, p in equation (7) should be replaced

by p', the fraction of Ri + Rtc inserted by Si. Similarly

if the potentiometer resistance Rpis significant, p

in equation (14) should be replaced by p" , the fraction

of R\ + Rp inserted by Si.

(Paper 75C3&4-321)

* with a high impedance detector. With

a galvanometer the formula for 6^ is

too complicated to he useful.

691
JOURNAL OF RESEARCH of the Notional Bureau of Standards — C. Engineering and Instrumentation

Vol. 73C, Nos. 3 & 4, July-December 1969 Design Features of a Precision AC-DC Converter

Louis A. Marietta

Institute for Applied Technology, National Bureau of Standards, Washington, D.C. 20234

and

Donald R. Flach

Institute for Basic Standards, National Bureau of Standards, Washington, D.C. 20234

(August 5, 1969)

With the availability today of high performance operational amplifiers and related components, it

is possible to construct an instrument for ac-dc transfer work that meets the requirements of a standards

laboratory. Transformation of measured average a-c values into d-c voltage is possible with a predict- able accuracy of 20 parts per million up to 1 kHz. Precision of operation is assured from dc to 100 kHz.

Key words: AC: DC: operational: precision: rectifier: transfer.

1. Introduction

Reference to precise a-c measurement has always

implied an rms evaluation. This has been so because

the majority of physical results in experiments and

other usage are related to transformation of electrical energy, to heat or mechanical work, and electrical laboratories have found the best accuracy in square- law responding instruments. With the availability

today of highly stable, nearly pure sine-wave power

sources and precision linear operational devices, the

situation has changed. For some ac-dc transfer work

such as amplitude monitoring [l]. 1 the average a-c responding systems to be described can have a su- periority of precision over rms converters. In addition,

the wide usage of average responding instruments

such as digital voltmeters with a-c options and sampling

devices has increased the need for a basic average

transfer standard. This paper will delve into design

factors that demonstrate the ability of transforming the average value into d-c voltage with a high order of accuracy.

2. Operational Rectifier Stage

The choice of systems and circuits for average a-c measurement is more limited than for rms. An average

1 Figures in brackets indicate the literature references at the end nf this paper.

responding circuit is involved with some form of

rectification — that is. it has to average either the positive or the negative values of the voltage wave, or both

without regard to sign. It is the defects in the rectifying elements that affect precision and accuracy. The

object of the operational rectifier circuit, figure 1. is to place the rectifying components in the feedback

loop at the output of the amplifier wherein the full loop gain amplification can be used to greatly reduce

the diode weaknesses. Th^ device is a unity gain cir- cuit (if equal value resistors are used at the input and

feedback connections) whereby half-sine waves are generated at each output.

Because of the unwieldy nature of transfer expressions in circuits with diodes, a linear amplifier type

analysis in the frequency domain will be pursued

first, by neglecting the diode imperfections.

2.1. Finite Gain Error

Using superposition theory and writing a current balance at the input summing junction, we find that the transfer function of a unity-gain inverting operational amplifier with perfect ratio resistors is v"JV]n= ~TTi2jAj)

il]

where .4/ is the open-loop voltage amplification of an

692
Fk.L'RE 1. Basic operational rectifier.

operational amplifier with ideal characteristics but

finite amplification.

To avoid commonly observed errors in the usual

exact-gain formula due to neglect of the phasor nature

of the open-loop voltage amplification. Ay. the general

formula can be expressed in a more usable form as follows v0jvm =-{zfiz>)

I (2 ) 1 + (llAf) (cos d-j sin0)[l + {ZfIZi)]

K

where 6 is the open loop phase angle of the amplifier,

and A/ is now the magnitude of the open loop amplification. Starting with zero degrees at direct current. 6 usually increases with a minus sign (lag) as the frequency increases. Zj and Z, are feedback and

input impedances, respectively. In most literature

a reader tends to interpret the closed-loop gain

expression as an output that falls off with frequency

because of the drop in open loop amplification.

Careful tests corroborate expression (2) and show

that this is not necessarily so. The output can actually

rise with frequency for a constant input in spite of

a dropping Af in the region where d has shifted beyond

90 degrees lag. the reason being that the phase lag

has created a real component of positive feedback.

The transfer error in ppm (parts per million) for an inverting unity gain situation is the difference

between unity and the magnitude of V, m/Vin . Except

for angles near 90°. the error can be closely approximated as ±{21Aj) • (cos d) • 106

: the minus sign

for phase lag less than 90° and plus sign for phase

lag greater than 90°. where {Aj > 1). Exactly at 90°

the error reduces to 2/Aj . and reaches zero error

slightly beyond 90°. However, effectiveness of quadrature error reduction does drop off rapidly for phase

deviations from the 90° region.

Since phase characteristics are rarely given in operational amplifier literature, the transfer error calculation of the operational rectifier will be hampered. Using the worst case basis of {2/Aj) • 10" for the error, for an amplifier with zero phase shift, it could be expected that an operational amplifier would

exhibit a gain transfer error of 100 ppm at the fre- quency where the open loop amplification is 20 000

and that the error would double each octave increase

of frequency. However, it was found from actual

test data that the error function did not behave in this manner but that the output remains quite constant

for a wide frequency range, in some cases rising at the higher frequencies due to the previously described

phase properties. Hence it can be assumed that the

(2/Aj) 10H error factor is quite conservative.

2.2. Finite Diode Conduction Level Error

Another source of error arises from the finite con- duction level of the diodes. Time is lost at the amplifier

in order for its output to reach the diode conduction

voltage, which means that there is a "lost area" at the beginning of the half sine wave output. The overall error in average value with a first order approximation can be expressed as the ratio of the lost small area to the half-wave area. It is desirable to analyze the factors

that may influence this particular error. For the period of a half-sine wave with Vm peak

volts, the average value is (2/7r)F,H . Hence the area under the function is (2Itt)(V„i) (rr) or 2Vm volt-radians.

The lost area in question can be found by formal

integration as

r<bc Lost Area.= V„, J sin 4>d<t> (4)

where </> r is the angle measured from the zero crossing

point to the diode conduction point. For small angles, cos $ = 1 — (<£

2 /2) and the lost area from expression

(4) is

Lost Area=V„,\—cos <$> |*<-=F„,(<^?/2) volt-radians. (5)

At angle (/> r the diode conduction voltage, Vc , at the

amplifier output is Vc = AjVs (6)

where Vx is the potential at the operational amplifier's

summing input junction and A/ is the open-loop voltage

amplification. Also for small radian angles

V, = 4>rVin . (7)

Substituting (7) into (6). the angle at conduction

point becomes

(f>r = Vr/VmAf. (8)

Using expressions (5) and (8)

Lost Area =ni2V„J$. (9)

Taking the ratio of the lost area (9) to the half-wave

area {2V,„). the overall error is

error. e=Vy4V}„A*. (10)

Because of the squared value for Ay in the expression.

693
the error due to the finite diode conduction level is considerably less than the closed-loop gain error noted

in expression (2).

2.3. Slew Rate Error A second source of possible error from the diode

conduction level is found in the consideration of the

operational amplifier — rectifier in the time domain.

Included in the data on operational amplifiers is the "slew rate" — a measure of the device's transient

response. Slewing rate.S. or maximum output voltage

change with respect to time is expressed as S= (dVu/dt ) mHX = ^(aM/lO6

) volts per microsecond

(11)

where V,, is the peak output voltage and oji, is the maximum frequency in radians per second for full output

of the device. Using the stated slew rate, the time

lost in reaching diode conduction can be found and

used to compute the lost area. The elapsed time

(tc ) from zero crossing to conduction. Vc is tc=VdS seconds. (12)

The angle. of expression (4) can be expressed as

4>c=(tjtn) (27T) radians (13)

where to is the time for one period of the operating

frequency. /o. Substituting (12) in (13)

<j) c = Vc 2irfJS radians. (14)

Using the lost area expression (5) and expression (14) for (t> c

Lost area due to S = Vm [Vc 2Trf»IS] 2 /2. (15)

As in expression (10) the overall error, e. comes from

the ratio of lost area to half-wave area.

error. e=F2(277/„) 2 /S 2

. (16)

Comparing expressions (16) and (10) — for example: if /o=16 kHz. Vc is about 0.6 V, an amplifier with

A/= 104 and a slew rate of 10 V per microsecond, the error from expression (10) with an input signal of Vm — 1 V becomes approximately two parts in 10".

Using expression (16) the error is about ten parts

in 106

. In this example the slew rate factor contributed

a greater error than the Af gain factor. However,

the slew rate figure alone is not a reliable measure

of an amplifier's ability to be used as a precision

operational rectifier. It is possible for a device to have

both a relatively low Ay and a high slew rate since

the two are independent design features of the amplifier. Such a rectifier would suffer a possible error from expression (10). Fortunately all amplifiers

investigated that have the desired open-loop amplification also have sufficient slew rate. One additional note is that amplifier output overshoot at the beginning and end of the half-sine wave has not been a troublesome factor. Thus far the discussion of the operational rectifier has been concerned with its a-c requirements: however

the precision of its average value output also demands

a strict dc stability. Not very long ago it was possible

to achieve low drift results only with chopper-stabilized devices. High a-c performance operational

amplifiers are now available in differential, nonchopper

design wherein the drift error contribution with a

5 \ input signal can be kept below 20 ppm for hours.

Bias current variations at the input are low enough

so as to be negligible in producing d-c output varia- tions with feedback resistors encountered (1 kfi

to 100 kfl).

2.4. Feed-Forward Operational Amplifiers

In the course of experimenting with a number of available operational amplifiers a certain group of them were found to exhibit a marked anomalous

behavior. These particular amplifiers showed a monotonically decreasing response with frequency

that was much sharper than was predictable with

the conservative error factor bases. An inquiry into design of these amplifiers revealed them as "feedforward" type devices. Briefly this means the input

signal assumes different paths through the overall amplifier chain depending upon the input frequency,

in the manner of a chopper stabilized amplifier.

They can sometimes be spotted in the literature

as wide-band devices with a high capacitance at the inverting input, as much as 0.02 jx¥. While feed- forward type operational amplifiers can be used as operational rectifiers. FET differential amplifiers

characterized by their low input capacitance at both inputs (5 pF or less) are generally preferred.

3. Operational Filter Stage

The operational stage following the rectifier is the

filter. (Fig. 2) Basically it is an accurate unity gain

stage with a large enough time constant to filter out the half-sine-wave ripple to a level tolerable to the d-c detector. It also provides nearly ideal impedance buffering between the rectifier output and the

d-c detector. The filter operational amplifier a-c response performance is unimportant, but like the

operational rectifier, the need for low d-c voltage drift must be met. To preserve the unity gain accuracy

and to present a low output impedance, the open loop

d-c voltage amplification should be at least 200 000 —

figure that is easily found in today's devices.

The feedback filter capacitor should have enough

capacitance to achieve the desired time constant in conjunction with the feedback resistor ( 10 kf) to 50 kH)

to meet the detector requirements. Capacitor leakage

resistance is important: the value should be at least 105 times the feedback resistor value. Polycarbonate

dielectric capacitors are available in a 40 ^F value,

meeting the low leakage requirement.-.

694
Fit. IRE 2. Basic operational filler.

4. Choice of Precision Resistors for the Rectifier

and Filter Stages

The principal factor in the gain-setting accuracy of an operational stage is the ratio accuracy of the circuit resistors external to the component amplifier. Wirewound resistors of various forms are available with

both individual and ratio accuracies of 5 ppm.

Extensive tests have demonstrated their ability to perform well in operational circuits. With little complexity it is possible to detect by d-c null tech- niques a difference of less than 5 ppm in a pair of high

quality wire wound resistors by interchanging their respective positions at the input and feedback of an

operational amplifier. The problem with some wire

wound resistors is that they can exhibit a noticeable

dc-ac error for frequencies above 5 kHz. amounting

to several hundred ppm at 50 kHz. While it is possible

to partially compensate for the error by using networks

tailored to the resistor characteristics, it is the purpose

of this article to describe systems of high precision

that are void of compensation schemes. Fortunately

there are available metal film resistors witli desirable

properties for the immediate application. These

possess a virtually zero dc-ac difference over the range

of interest (dc to 100 kHz). They are readily available

with better than 100 ppm ratio accuracy with a relative temperature coefficient of a few ppm per degree. For

wide band work in the area of 100 ppm accuracy, the metal film resistors are highly recommended: for high-accuracy, low-frequency applications of say 20

ppm. wire wound resistors may be more suitable.

5. Selection of Diodes at Operational Rectifier

The role of reverse leakage current and junction

capacitance of a chosen diode must be considered in relation to the feedback resistor. It is possible to use

a single silicon diode in each feedback path of the

rectifier circuit since they are available with leakage

currents less than 10~9 A. The usual operational for- ward diode current is 10_:5 A. hence the error is 1 ppm

for the stated leakage current. A more conservative

approach is to use a pair of diodes in series, with a

resistive shunt path between them for the leakage

current: an arrangement conceived by Richman [1]. The series feature also reduces the resultant capaci- tance at full reverse bias: typically 1 pF.

The junction capacitance at the diode can introduce an error by adding a component of reverse current to the diode's forward signal current. However the net

effect can be small because of the quadrature phase

relationship of the currents. The error expression can be derived from the circuit impedances in the current branches. For small values of capacitance or high ratio of capacitive reactance to series feedback

resistance, the error becomes = 1/2 {RjjX,-f 10fi ppm.

For example at an operating frequency of 16 kHz and

a feedback resistance of 104 Q. a 1 pF diode capaci- tance would introduce an error of about 1/2 ppm.

With two silicon diodes in series the output must

reach about 1.2 V before conduction takes place.

This quadruples the small error in average value be- cause of the "lost area" as explained previously. The

latest development of "hot carrier" diodes can be used

to good advantage. These high-speed diodes have the

low-leakage feature of silicon diodes and a low forward

conduction voltage similar to germanium less than

0.39 V. Reverse recovery or effective minority carrier life- time in a diode is a measure of its ability to turn off

in going from forward to reverse conduction. In the

operational rectifier this factor affects the termination

of the half sine wave function, resulting in an error in average value. The analysis of the error is nearly the same as for the case of limited amplifier slew rate dis- cussed previously with the error showing up as a lost area. Fortunately the diodes described in this section

have recovery times in the vicinity of 10~ 9 s. Using

the same procedure as with the example of the slew

rate discussion, the error contributed at 16 kHz with

the stated recovery time is found from expression (6). The error is equal to 0

2 /4 or (105

• K)-9 r74= 10~8/4

which is negligible.

6. Stabilization of Operational Rectifier

To prevent high frequency oscillation a small

capacitor is generally needed across the feedback

resistor of an operational device. The resultant time

constant places an upper corner frequency. Fi,. having a direct bearing on the gain accuracy at some

lower operating frequency. F». For large separation

of the two frequencies, the asymptotic slope error

is nearly (Fu/F),)2 - 10H. in ppm where Fu = 1/2 nRjCj.

The analysis is correct for a linear system: however

in the rectifier configuration the capacitor is across both the feedback resistor and diode string resulting

in a marked reduction of corner frequency. Fi, which

is difficult to calculate. Experience has shown that a series pair of 5 pF capacitors has been effective

in stabilizing all amplifiers tested. However, this value

of stabilizing capacitance can degrade the gain

accuracy by 50 ppm at 20 kHz with a 5 kfi feedback

resistance.

7. Unity-Gain Input Isolation Amplifier Stage

The combination of the operational rectifier stage

and filter stage is sufficient to meet the needs ol most

695
average ac-dc transfer requirements. This system

has an input impedance equal to the input resistor of the operational rectifier stage, typically 5 ktl. In some cases this may be annoying because of the

loading affect on a-c sources and a-c attenuators. For these reasons the effectiveness of an operational

unity-gain impedance transforming circuit was

investigated for use in applications where 100 ppm

accuracy is sufficient. The basic circuit is show n in figure 3. It is recognized

as a noninverting amplifier with unity gain for the indicated input-output connection. Note that no

precision resistors are involved in its operation and

that it is able to take advantage of the high input

impedance available with FET type differential devices — around 10" fi shunted by 3.5 pF of capacitance. In order for the stage to have the desired gain

accuracy from d-c through the highest operating

frequency, the selected operational differential amplifier must have both a wide-band high open

loop amplification and a wide-band high common-mode

rejection ratio. CMRR. The latter factor is a measure

of the amplifier's ability to precisely respond only

to the instantaneous amplitude difference value

between the two input terminals: hence it desirably

produces no output when both inputs are at the same

potential and polarity. Analytically the CMRR of a differential amplifier is the ratio of a voltage V m

to the difference of the potentials at the two inputs

required for a null output; expressed as CMRR=V„l[(Vll ±e+ )-(Vll ±e-)]. (17)

Each of the terms in the denominator of (17) is an

equivalent potential at each input consisting of a magnitude VH and an apparent error potential. ±e.

For example if V„ is 1 V and the difference of the two equivalent input potentials is 0.001 V. the CMRR

is 1000 : 1. Its polarity is a function of the polarity of the symmetry inperfections. and e_. Writing

circuit voltage loops, the transfer function of the

stage is expressed as

V IV. =

1±1/CMRR

,18) ' out/ 'in -I,/ n -• a \ I A 1101

1 + ( cos 0—j sin u)IAf

The effect of wide band CMRR and .-/, with phase

lag that is greater than 90°. can be seen in figure 4 for three situations: (A) high CMRR and low A/: (B) high

A, and low CMRR: (C) both CMRR and A, are high.

The figure is a plot from laboratory data on actual

devices and corroborates and predicted behavior

Fn.LRE 3. I nity-gain isolation amplifier.

from expression (18) though the slope of curve (B)

could have been lower for a minus polarity effect of the CMRR. The three devices shown in figure 4 are high-performance units, and the use of the terms

"low A/' or "low CMRR are in a relative sense, keeping

in mind that the total ordinate value is less than 0.1 percent error. The stage also has a desirable low output imped- ance—a dynamic impedance equal to the amplifier

intrinsic output impedance divided by the value of

A,, A typical amplifier is capable of delivering ±10

V output at 10* Hz. with a load current of 10 mA. In summary, the operational stage described is a rather

elegant impedance transformer with an impedance

ratio of at least 106

: 1 operating from dc through

100 kHz with a irain accuracv of 200 ppm to at least 20 kHz.

8. Test Results

A large number of tests were performed involving

a variety of operational amplifiers for use as precision

rectifier, filter, and unity-gain impedance coupling

stages. The tests could be grouped in three broad

areas of interest: accuracy at a reference frequency,

precision (stability and repeatability) and frequencyresponse (ac-dc deviation).

8.1 . Tests for Accuracy

The first group of tests were concerned with the

accuracy of transformation of ac to dc by an operational rectifier: i.e.. the relation between the average

value of the output (half-wave) and the rms value of the input, with a voltage wave of very low distortion, so that V.dve = VTms • V2/7T. The operational amplifiers available for these tests were of the chopperstabilized type having very high open-loop amplification at frequencies below 5 kHz. Their d-c drif. was less than 5 fx\ for periods of several hours. Oilfilled, wire-wound resistors matched to a few ppm in unity-ratio value were employed, with resistance

values from 1 to 25 kfi. The major components of the measuring system for these tests, shown in figure 5. included a low-distortion, highly stable a-c power

source, (total harmonic distortion below 0.01%) a seven-dial calibrated, precision d-c power supply,

a standards laboratory differential thermal voltage

comparator (called a DTYC). [4] sensitive d-c null meter, and the average ac-dc converter under test. The magnitude of each harmonic of the source (up

through the 9th) was measured. Its effect on the form

factor of the voltage wave was calculated to be less than 10 ppm. assuming the worst possible phase angle

with respect to the fundamental.

The standard was the rms responding DT\ C unit,

having the function of monitoring the sine-wave signal

to be measured and matching it by null method to the

d-c supply of the equivalent rms value. In turn the , d-c supply was scaled down by a factor of V2/tj\ by

the attenuator, and nulled against the d-c output of the average converter. The d-c difference is the error of

696
PRECISION AC

SOURCE

5V. RKS

AVERAGE CONVERTER UNDER

TEST

PRECISION OC

SOURCE 5¥. OC L ATTENUATOR

AC-DC

DIFFERENCE

vT

FIGURE 5. Testing system.

the device under test. Since the entire system has d-c response, scale factor inaccuracies at any position in the system could be checked by substituting the same d-c supply potential for the a-c signal (the average

unit has identical input-output polarity correspondence). By using the same d-c supply as both signal

and reference, its amplitude could be varied at will, allowing a check on dynamic range and on d-c offset. A 5 V rms a-c level was used in all the tests in order

to avoid approaching the usual ± 10 \ operational

amplifier output limit. Operating voltages below 5 V

are not recommended because most of the error factors discussed would be increased.

The results of these tests in figure 6 show that the

operational rectification and filtering combination could

yield a d-c voltage whose amplitude can be related to the rms value of a-c input with an accuracy of better than 20 ppm for frequencies out to 1000 Hz. At higher

frequencies the accuracy was degraded because this was a feed forward type of amplifier. Short term perturbations of output were in the vicinity of 1 ppm. and

possibly less. These tests have demonstrated high accuracy

performance at low and mid range audio frequencies.

The next emphasis was on the design factors needed to extend the operating frequency range to 100 000 Hz

and to include a high input impedance stage ahead of the rectifier.

8.2 Tests for Frequency Response

This group of tests involved the latest differential FET operational amplifiers having wide-band, highvalue open-loop amplification figures. In addition,

for reasons previously stated in the section on precision

resistors, metal film resistance components were used

in these tests. Since the question of measurement

accuracy had been settled in the previously described

group of tests, the test circuit arrangement was

modified to demonstrate operational precision. Rather

than employing the primary standard DTVC unit, a much simpler version of differential thermal comparator was used to monitor the output of the a-c signal power source for the purpose of maintaining

constant amplitude. A peak ac-dc voltage comparator,

though sensitive to waveform distortion, could be

used in place of an rms comparator since an ac source of low distortion is specified. An in-depth

analysis of the effects of ac distortion on the rectified average value can be found in reference [3]. For these

tests the stress was on frequency response, exact value of magnitudes being of secondary importance.

The recorded values were the measured differences

697
00 400 1000 ?5O0 4800 10000

FREQUENCY Hi

Kl«.i RK f>. Transfer tircurary nj uc-ilr converter iC.hiipper-slahilizetl amplifier/.

between the average d-e output and the fixed d-c reference pnuer supply, expressed in ppm and plotted versus frequency. Normalizing the error values at some low frequency starting point is a means ol removing: the error offset due to any imprecision ol resistor ratio and d-c offset at the amplifier stages.

The crosses in figure 7 show the test values obtained

with an operational rectifier-filter combination which

had metal film resistors.- Considering the number ol instruments involved in the tests and their individual

contribution to the end precision, the assignment ol' ± 150 ppm for the wide-band ac-dc error of the average

measuring instrument is conservative. At the same

time of' the tests shown in figure 7. a unity-gain input

isolation amplifier stage was added ahead of the operational rectifier state. In figure 7. the circles show

the results of three runs: again all the values taken

are plotted. Comparison ol the circles and crosses shows that very little was changed in the response

characteristic with the addition of the isolating

amplifier. The benefit realized is substantial: the impedance presented to the a-c source increased from

5 kfl to 10 \1Q. Ml frequency test points were within

200 ppm of the normalized value.

9. Some Wiring Suggestions

Circuit wiring, especially in the operational rectifier circuit should be short for minimum lead inductance

and arranged in a manner for minimum shunt capacitance. One of the most critical points in the connection

from the selected polarity output of the rectifier to the input resistor of the filter stage. Distortion of the

half-sine wave resulting from shunt capacitance at this point could affect the accuracy of the average

value.

K<iur wrrr maile ji Ka<'h freiiupnry. bui vw-rr rn<i .i-[urji*-K |ilitlli'ij hImmi iJir faults <]rflVrp«i bv itnK Mnall amounts.

It is strongly suggested the electronic circuit be

d-c isolated from chassis ground. Removing the input

and output common terminal from ground helps in eliminating the noisy effects of circulating d-c ground

currents. Both a-c and d-c isolation can be had by a guarded circuit which offers low electrical capacity

from the common terminals to chassis. This form of

isolation requires a box-w ithin-a-box type construction

plus a special power supply. Such regulated supplies

with a multi-shielded transformer are available with

10 000 megohm d-c insulation and 0.1 pF line-power

input to d-c output isolation.

10. Initial D-C Offset Voltage Adjustments

Alter a sufficient warm-up period (usually 1 hr) the voltage offset of the filter stage is trimmed for zero out- put voltage with its input connection from the input

resistor momentarily tied to common. The operational

rectifier stage offset can be likewise adjusted by

shorting to common the input side of the input resistor and trimming for zero output voltage. For this stage the diodes must be temporarily short-circuited. Effective- ness of input-output transfer of the instrument can be

observed by applying some d-c value from 1 to 10 V

to the input and comparing it to the instrument output

by means of a null meter: the diode rectifier polarity

output must be correctly selected. The d-c inputoutput agreement should be within the ratio accuracy

of the resistors at the rectifier and filter stages.

1 1. Suggested Circuit

Figure 8 is a schematic diagram ol a line-up ol operational stages for use as an average ac-dc con- verting instrument. The circuit is capable ol serving

either the need for accuracv or tor operational

precision.

698
1 - A RUNS WITH 5kH 1MPUT AT OPERATIONAL RECTIFIER

° - 3 RUNS fITH A UNITY CAIN AMPLIFIER, lONfl INPUT, AHEAO

OF OPERATIONAL RECTIFIER

FREOUENCT Hi

FlOURE 7. Frequency response of conierte r

HIGH INPUT IMPEDANCE

UNITY GAIN AMP

HALF IAVE RECTIFIER FILTER

WMF - LOW LEAKAGE

ALL RESISTORS WITH P SUFFIX ARE

PRECISION TYPE Fid RE 8. Schematic diasrum.

If accuracy of 20 ppm is desired for ac-dc transfer,

wherein the measured values can be related on an rms

basis, the following points are pertinent. The high input

impedance operational first stage must be deleted and

the input signal applied to the second stage — the operational rectifier. The reason for this precaution is that the available amplifiers for use as a noninverting

stage have marginal open loop gain and CMRR

figures for accuracy work below 100 ppm. As for the

rectifier stage, the selected operational amplifier must

have a very high open loop gain and a low d-c drift. An open loop gain of 200 000 at 1 kHz is needed if the contributed error is to be less than 10 ppm. The

voltage drift should be in the area of 1 /xV/°C and

10- 1U A/°C for the current drift. While differential FET type operational amplifiers come close to meeting

these specifications, the accuracy results reported in

this writing have been obtained with chopper stabilized

devices. Because of their possible feed-forward design

the 20 ppm operating accuracy is limited to frequencies

below 1 kHz and the gain setting resistors at the

rectifier stage are restricted to values of 1 kfi or less. The latter components should have a ratio accuracy

error of less than 5 ppm. available in oil filled, wire

wound resistors. A signal level of 5 \ rms at the a-c source is suggested. The a-c signal power source

should have a total harmonic distortion of less than 50

ppm and freedom from a d-c component, if the rms

value is desired. The operational amplifier at the filter stage need only have the stated d-c precision of the

rectifier stage.

The three stages shown in figure 8 can offer an operating precision and flatness of amplitude response

with frequency of better than 200 ppm over the range

from d-c to 100 KHz. as described previously under

the section heading of Test Results. FET-type differ699
ential operational amplifiers can be found to meet the

requirements of each stage. If the sole interest is operational precision in the form of stability and repeatability, the principal concern in selecting the operational amplifiers is the d-c drift characteristics.

However, the open loop voltage gain has to be con- sidered in order to maintain the desired transfer gain

ratio. With a voltage drift value below 10 fx\ l°C and a current drift under 10~ 1M A/oC the operating precision can be held to better than 100 ppm. For situations re- quiring accuracy of measurement or if flatness of amplitude response with frequency is important, the

wide-band open loop voltage gain deserves additional

attention. In the case of the noninverting first stage

the CMRR figure is also important. If the input

stage is to have a unity gain accuracy of 100 ppm at 5 kHz. both the open loop gain and the CMRR at that frequency should be at least 20 000 for the selected

operational amplifier. For the rectifier stage the same

values of 20 000 is needed at 5 kHz for 100 ppm

accuracy but the CMRR can be neglected. The filter

stage as usual need only have the stated d-c operating

characteristics. Again, in the case of nns measure- ments ot accuracy the a-c signal source must have

appropriate purity — no more than 200 ppm total harmonic distortion for 100 ppm transfer accuracy.

12. References

[1] Richman. P.. Operational Rectifier. U.S. Patent No. 3311835. March 28. 1967. assigned to Weston Instruments. Inc. [2] Marzetta. L A.. \ Peak AC-DC Voltage Comparator for Use

in a Standards Laboratory. NBS Technical Note No. 280. issued January 17. 1966.

[3] Richman. P.. Modern AC Voltage Calibration for Audio and

Sub- Audio Frequencies, presented at 19th Annual ISA Conference. October 12-15. 1964. New York. Preprint No. 21. 4-2-64.

[4] Hermach. F. L.. Criffin. J. E.. and Williams. E. S., A System for Accurate Direct & Alternating Voltage Measurements. IEEE Trans, on Instrumentation & Measurement IM14

p2 15-224. 1965.

(Paper 73C3&4-288)

700
APPENDIX I

Electrical and Electronic Metrology: A Bibliography

of NBS Electrosystems Division's Publications

(January 1968 through February 1984) 701

APPENDIX II

Metrology for Electromagnetic Technology:

A Bibliography of NBS Publications

(January 1970 through December 1983) 732

APPENDIX III

A Bibliography of the NBS Electromagnetic Fields

Division Publications

(January 1982 through December 1983) 795

B Bibliography of the NBS Electromagnetic Fields

Division Publications

(January 1980 through December 1981) 808

C A Bibliography of Publications in the NBS Electromagnetic

Fields Division

(January 1970 through December 1979) . . . 821

701
APPENDIX I

Electrical and Electronic Metrology:

A Bibliography of NBS Electrosystems Division's Publications

Compiled by

Jenilie C. Palla and Betty A. Oravec

Edited by

John R. Sorrel Is

702
Abstract

This bibliography covers publications of the

Electrosystems Division, Center for Electronics and

Electrical Engineering, NBS, and of its predecessor

sections for the period January 1968 to March 1984.

A brief description of the Division's technical

program is given in the introduction.

Introduction

Scope and Organization

This bibliography covers publications by the Electrosystems Division, CEEE,

plus those of the predecessor sections for the Electricity Division beginning

with 1968. For convenience, the references are grouped into ten categories by

subject: DC and Low Frequency Measurements, Thermal Converters, Power Systems

Related Measurements, High Voltage Measurements, Data Acquisition and

Conversion, Automatic Testing, Electrical Insulation Measurements, Electronic

Instrumentation and Metrology, Sensors, and Miscellaneous.* References may

appear in more than one category; the reader is urged to check several

categories if an overlap is likely. References are presented chronologically

within each category, with the most recent last. An index by author is also

provided.

Division Activities

The Electrosystems Division develops measurement methods and standards and

provides metrology support for electrical and electronic systems, components

and materials; it develops and maintains national reference standards for

practical electrical units derived from basic units. The measurement

technology developed by the Division is applied to the advancement of modern

electrical /electronic measurement instrumentation, to the development of

efficient generation, transmission and utilization of electric energy, and to

maintaining a national reference base for certain critical measurements

required by science, industry, amd government agencies.**

Presently, the Division is structured into two groups — (1) Applied

Electrical Measurements, and (2) Electronic Instrumentation and Metrology.

*Responsibi 1 i ty for work on sensors (transducers) for physical quantities

such as pressure and acceleration was reassigned outside the Center several

years ago; reference to a selected short list of papers is provided for

completeness.

**Research, development and calibration work on basic electrical units

is conducted by the Electrical Measurements and Standards Division of

NBS. For additional references the reader is referred to the NBS

Publication LP- 38 Electrical Units, Instruments, and Measurements,

available from the Electricity Division, Metrology Building, Room B258

,

National Bureau of Standards, Washington, DC 20234.

703
Applied Electrical Measurements

Within the Applied Electrical Measurements (AEM) Group, standards are developed and calibrations performed for electric power and energy,

high-voltage and high-power impedances, instrument transformers, high-voltage

dividers, and high-voltage transient measurement systems utilized in

measuring such phenomena as standard lightning impulses and voltage/current

pulses encountered in x-ray equipment. In situations where high-voltage

measuring instruments must be calibrated in their working environment or are too bulky for shipment to NBS, calibrations on a limited basis are performed on the customers' premises.

The AEM Group is also concerned with the measurement of electric fields

and ion activity in the vicinity of high-voltage ac and dc transmission

lines. Measurement methods to monitor or predict insulation failure

mechanisms are developed with emphasis on liquid, liquid-solid, and

gaseous insulation systems.

Electronic Instrumentation and Metrology

The Electronic Instrumentation and Metrology (EIM) Group is concerned

with modern electronic techniques for the measurement of electrical

parameters in the dc and low-frequency (to about 1 MHz) range. Among

the approaches investigated, very prominent are dynamic measurement

methods in which electrical signals are measured by fast sampling

techniques. Data from the measurements is processed digitally utilizing

mi croprocessor-based i nstrumentation.

Within the EIM Group, methods are developed to characterize data

acquisition and conversion devices. A calibration service for

precision A/D and D/A converters has recently been offered and phase

angle calibration based on a dual sine-wave source is now available

on special request. Research is continuing on new precision signal sources based on digital synthesis. Such signal sources are expected

to serve as calibration standards for numerous parameters. Special

purpose instrument standards, e.g., wide-band sampling wattmeters, are under development, and automation techniques for precision measurement

are being explored in the area of alternating voltages.

704
How To Obtain Listed Publications

In most cases reprints of articles in technical journals may be

obtained on request to the author or to the Electrosystems Division,

Room B166, Metrology Building, National Bureau of Standards,

Washington, DC' 20234.

Some of the publications are available from the Government Printing

Office (GPO). Current price information may be obtained from GPO,

Order Inquiry Section, by calling 202/783-3238. Remittances for

publications to be purchased from GPO should be mailed to

Superintendent of Documents, U.S. Government Printing Office,

Washington, DC 20402, by postal money order, express money order,

or check. Postage stamps will not be accepted. Foreign remittances

should be marte either by international money order or draft on an American bank. The Superintendent of Documents allows a discount

of 25 percent on orders of 100 or more copies of one publication.

The GPO stock number and full title of the publication must be given

when ordering. For credit card orders, information is needed as fol lows.

WS4*

Credit Card Order* Only

Total charges $ Fill In the boxes below.

Credit | Card No. I

Expiration Date Month/Year

Persons who make frequent purchases from GPO may find a deposit

account convenient. Deposits of $50 or more are accepted against

which orders may be placed without making individual remittances or first obtaining quotations. Order blanks are furnished for this

purpose. After the order has been processed, the order itself is

returned, showing the publications supplied, explanations regarding

those not sent, the amount of charge, and the balance on deposit.

705
Copies of most NBS publications, NBS Interagency Reports (NBSIR

Series), and other Government reports may be ordered through the

National Technical Information Service, U.S. Department of

Commerce, Springfield, VA 22161. Current price information may be

obtained from the NTIS Order Desk by calling 703/487-4650. If

there is an option as to form, be sure to specify whether paper

copy (PC) or microfiche (MF) is desired. As long as the

publication is available in paper copy from GPO, only microfiche

copy can be obtained from NTIS. Prepaying (with check or money

order) or using an NTIS Deposit Account or American Express Card

Account speeds order processing. Be sure payment and order are

sent together. Checks, payable to NTIS, must be in U.S. dollars.

For convenience, a table providing GPO stock numbers and NTIS

accession numbers for all NBS publications included in this

bibliography is given on pages 28-30.

706
Bi bl i ography

DC and Low Frequency Measurements

W. C. Sze, "An Injection Method for Self-Calibration of Inductive

Voltage Dividers," J. Res., Nat. Bur. Stand. (U.S.), (Eng. & Instr.)

72C , No. 1, pp. 49-59 (Jan. -Mar. 1968).

0. Petersons and W. C. Sze, "Calibration of Transformer Ratios With

a Reciprocal Transformer-Type Network," Conf. Precision Electromagnetic

Measurements Digest, Boulder, CO, pp. 57-59, June 2-5, 1970.

R. F. Dziuba and B. L. Dunfee, "Resistive Voltage-Ratio Standard and

Measuring Circuit," IEEE Trans. Instrum. Meas. IM-19 , No. 4, pp. 266-277

(Nov. 1970).

A. Erez, "Low-Frequency Electrical Signal Measurement by Electrooptical

Methods," IEEE Trans. Instrum. Meas. IM-21 , No. 4, pp. 358-360 (Nov. 1972)

T. M. Souders, "Wide-Band Two-Stage Current Transformers of High Accuracy,

IEEE Trans. Instrum. Meas. IM-21 , No. 4, pp. 340-345 (Nov. 1972).

W. C. Sze and F. R. Kotter, "The Design of Near-Perfect Instrument

Transformers of Simple and Inexpensive Construction," J a Appl . Meas. 2,

pp. 22-27 (1974).

H. K. Schoenwetter, "An Ultra-Stable AC Power Supply for an Absolute

Volt Determination," Metrologia 10, pp. 11-15 (1974).

D. B. Sullivan and R. F. Dziuba, "Low Temperature Direct Current

Comparators," Rev. Sci. Instrum. 45_, No. 4, pp. 517-519 (Apr. 1974).

K. J. Lentner, "A Current Comparator System to Establish the Unit

of Electrical Energy of 60 Hz," IEEE Trans. Instrum. Meas. IM-23 , No. 4,

pp. 334-336 (Dec. 1974).

T. M. Souders, "An Audio-Frequency Four-Terminal Resistance Bridge,"

IEEE Trans. Instrum. Meas. IM-23 , No. 4, pp. 342-345 (Dec. 1974).

D. B. Sullivan and R. F. Dziuba, "A Low-Temperature Direct- Current

Comparator Bridge," IEEE Trans. Instrum. Meas. IM-23 , No. 4, pp. 256-260

(Dec. 1974).

R. S. Turgel, "Digital Wattmeter Using a Sampling Method," IEEE Trans.

Instrum. Meas. IM-23 , No. 4, pp. 337-341 (Dec. 1974).

R. F. Dziuba and D. B. Sullivan, "Cryogenic Direct Current Comparators

and Their Applications," IEEE Trans. Magn. MAG-11 , No. 2, pp. 716-719

(Mar. 1975).

R. S. Turgel, "Sampling Techniques for Electric Power Measurement," Nat.

Bur. Stand. (U.S.), Tech. Note 870, 35 pages (June 1975). [GP0]

707
R. S. Turgel , Chapter 9.5 Voltage, Current, and Charge Measurements,

Methods of Experimental Physics , Vol. 2, Electronic Methods, 2nd Edition,

Part B, Academic Press, Inc., 1975.

F. L. Hermach, "AC-DC Comparators for Audio-Frequency Current and Voltage

Measurements of High Accuracy," IEEE Trans. Instrum. Meas. IM-25 , No. 4,

pp. 489-494 (Dec. 1976).

C. B. Childers, R. F. Dziuba, and L. H. Lee, "A Resistive Ratio Standard for

Measuring Direct Voltages to 10 kV," IEEE Trans. Instrum. Meas. IM-25 , No. 4,

pp. 505-508 (Dec. 1976).

H. K. Schoenwetter, "An RMS Digital Voltmeter/Calibrator for Very-Low

Frequencies," IEEE Trans. Instrum. Meas. IM-27 , No. 3, pp. 259-268

(Sept. 1978).

N. B. Belecki, B. L. Dunfee, and 0. Petersons, "The National Measurement

System for Electricity," NBSIR 75-935, 86 pages (Sept. 1978).

R. S. Turgel and N. M. Oldham, "High-Precision Audio-Frequency Phase

Calibration Standard," IEEE Trans. Instrum. Meas. IM-27 , No. 4, pp. 460-464

(Dec. 1978).

B. F. Field, "A Fast Response Low-Frequency Voltmeter," IEEE Trans. Instrum.

Meas. IM-27 , No. 4, pp. 368-372 (Dec. 1978).

R. C. McAuliff, K. J. Lentner, W.J.M. Moore, and G. Schuster, "An

International Comparison of Power Measurements at 120 V, 5 A, and 60 Hz," IEEE

Trans. Instrum. Meas. IM-27 , No. 4, pp. 445-449 (Dec. 1978).

R. L. Kahler, "An Electronic Ratio Error Set for Current Transformer

Calibrations," IEEE Trans. Instrum. Meas. IM-28 , No. 2, pp. 162-164

(June 1979).

H. K. Schoenwetter, "NBS Provides Voltage Calibration Service in 0.1 - 10-Hz

Range Using AC Voltmeter/Calibrator," IEEE Trans. Instrum. Meas. IM-28 , No. 4,

pp. 327-331 (Dec. 1979).

K. J. Lentner, C. B. Childers, and S. G. Tremaine, "A Semiautomatic System for

AC/DC Difference Calibration," IEEE Trans. Instrum. Meas. IM-29 , No. 4,

pp. 400-405 (Dec. 1980).

B. A. Bell, B. F. Field, and T. H. Kibalo, "A Fast Response, Low-Frequency

Sampling Voltmeter," Nat. Bur. Stand. (U.S.), Tech. Note 1159, 113 pages

(Aug. 1982). [GP0]

N. M. Oldham, "A 50 ppm AC Reference Standard Which Spans 1 Hz to 50 kHz,"

IEEE Trans. Instrum. Meas. IM-32 , No. 1, pp. 176-179 (March 1983).

K. J. Lentner and D. R. Flach, "An Automatic System for AC/DC Calibration,"

IEEE Trans. Instrum. Meas. IM-32 , No. 1, pp. 51-56 (March 1983).

H. K. Schoenwetter, "AC Voltage Calibrations for the 0.1 Hz to 10 Hz Frequency

Range," Nat. Bur. Stand. (U.S.), Tech. Note 1182, 58 pages (Sept. 1983).

[GPO]

708
Thermal Converters

E. S. Williams, "A Voltage Converter for a New Era," Meas. Data

(Nov. /Dec. 1968).

L. A. Marzetta and D. R. Flach, "Design Features of a Precision

AC-DC Converter," J. Res., Nat. Bur. Stand. (U.S.), (Eng. & Instr.)

73C, Nos. 3 and 4, pp. 47-55 (July-Dec. 1969).

E. S. Williams, "Calibration of Thermal Voltage Converters," Proceedings,

Third Precision Measurements Association Conference, Burbank, CA (197U).

E. S. Williams, "Thermal Voltage Converters and Comparator for Very

Accurate AC Voltage Measurements," J. Res., Nat. Bur. Stand. (U.S.),

(Eng. & Instr.) 75C, Nos. 3. and 4, pp. 145-154 (July-Dec. 1971).

F. L. Hermach, "AC-DC Comparators for Audio-Frequency Current and

Voltage Measurements of High Accuracy," IEEE Trans. Instrum. Meas. IM-25 , No. 4, pp. 489-494 (Dec. 1976).

F. L. Hermach and D. R. Flach, "An Investigation of Multi junction Thermal

Converters," IEEE Trans. Instrum. Meas. IM-25 , No. 4, pp. 524-528

(Dec. 1976).

E. S. Williams, "Thermal Current Converters for Accurate AC Current

Measurement," IEEE Trans. Instrum. Meas. IM-25 , No. 4, pp. 519-523

(Dec. 1976).

K. J. Lentner and S. G. Tremaine, "A Semiautomatic AC/DC Thermal Voltage

Converter Calibration System," NBSIR 82-2576, 62 pages (Sept. 1982).

K. J. Lentner and D. R. Flach, "An Automatic System for AC/DC Calibration

IEEE Trans. Instrum. Meas. IM-32 , No. 1, pp. 51-56 (March 1983).

709
Power Systems Related Measurements

0. Petersons, "A Transformer-Ratio-Arm Bridge for Measuring Large

Capacitors Above 100 Volts," IEEE Trans. Power Appar. Syst. PAS -87 , No. 5, pp. 1354-1361 (May 1968).

0. Petersons, "A Semi -Automatic Bridge for Measuring Power Losses in

Shunt Reactors," Proceedings of 23rd Session International Conf. Large

High-Tension Electric Systems, Paris, France, pp. 6-8 (Aug. 24-Sept. 3, 1970).

T. M. Souders, "A Wide Range Current Comparator System for Calibrating

Current Transformers," IEEE Trans. Power Appar. Syst. PAS-90 , No. 1,

pp. 318-324 (Jan. /Feb. 1971).

T. M. Souders, "Wide-Band Two-Stage Current Transformers of High Accuracy,"

IEEE Trans. Inst rum. Meas. IM-21 , No. 4, pp. 340-345 (Nov. 1972).

D. L. Hillhouse and A. E. Peterson, "A 300-kV Compressed Gas Standard

Capacitor With Negligible Voltage Dependence," IEEE Trans. Instrum. Meas.

IM-22 , No. 4, pp. 408-416 (Dec. 1973).

K. J. Lentner, "A Current Comparator System to Establish the Unit of

Electrical Energy at 60 Hz," IEEE Trans. Instrum. Meas. IM-23 , No. 4,

pp. 334-336 (Dec. 1974).

R. S. Turgel , "Digital Wattmeter Using a Sampling Method," IEEE Trans.

Instrum. Meas. IM-23 , No. 4, pp. 337-341 (Dec. 1974).

R. S. Turgel, "Sampling Techniques for Electric Power Measurement,"

Nat. Bur. Stand. (U.S.), Tech. Note 870, 35 pages (June 1975).

S. R. Houghton, "Transfer of the Kilowatthour," IEEE Trans. Power Appar.

Syst. PAS-94 , No. 4, pp. 1232-1240 (July/Aug. 1975).

0. Petersons and W. E. Anderson, "A Wide-Range High-Voltage Capacitance

Bridge With One PPM Accuracy," IEEE Trans. Instrum. Meas. IM-24 , No. 4,

pp. 336-344 (Dec. 1975).

N. M. Oldham, "A Measurement Assurance Program for Electric Energy,"

Nat. Bur. Stand. (U.S.), Tech. Note 930, 23 pages (Sept. 1976).

[GP0]

F. R. Kotter and M. Misakian, "AC Transmission Line Field Measurements,"

prepared for the Department of Energy, DOE Report No. HCP/T-6010/E1,

pp. 1-79 (Nov. 1977).

D. L. Hillhouse and W. C. Sze, "Calibration of CCVTs in the Substation,"

45th Annual International Conference of Doble Clients, Section 9-501

(1978).

710
0. L. Hillhouse, 0. Petersons, and W. C. Sze, (Principal Investigators), "A

Prototype Field Calibration System for Coupling Capacitor Voltage Transformers

(CCVTs)," Final Report No. EL-690, prepared for the Electric Power Research

Institute, under Project RP-134-1 (Apr. 1978).

D. L. Hillhouse, "Portable System Calibrates CCVTs," Electr. World 189,

No. 12, pp. 44-46 (June 15, 1978).

W. E. Anderson and R. S. Davis, "Dissipation Factor Measurements on Dielectric

Materials at Liquid Helium Temperatures," 1975 Annual Report of the Conference

on Electrical Insulation and Dielectric Phenomena, National Academy of

Sciences, Washington, DC, pp. 151-156 (July 1978).

W. E. Anderson, R. S. Davis, 0. Petersons, and W.J.M. Moore, "An International

Comparison of High Voltage Capacitor Calibrations," IEEE Trans. Power Appar.

Syst. PAS-97 , No. 4, pp. 1217-1223 (July/August 1978).

Working Group on Electrostatic and Electromagnetic Effects of Transmission

Lines, "Measurement of Electric and Magnetic Fields From Alternating Current

Power Lines," (NBS participants: F. R. Kotter and M. Misakian), IEEE Trans.

Appar. Syst. PAS-97 , No. 4, pp. 1104-1114 (July/August 1978).

M. Misakian and F. R. Kotter, Discussion of Paper F78 169-5, "Analysis of the

Proximity Effects in Electric Field Measurements," IEEE Trans. Power Appar.

Syst. PAS-97 , No. 6, pp. 2175-2176 (Nov. /Dec. 1978).

R. C. McAuliff, K. J. Lentner, W.J.M. Moore, and G. Schuster, "An

International Comparison of Power Measurements at 120 V, 5 A, and 60 Hz," IEEE

Trans. Instrum. Meas. IM-27 , No. 4, pp. 445-449 (Dec. 1978).

M. Misakian, "Measurement of AC Transmission Line Fields and Related

Electrical Parameters," IEEE Tutorial Text 79-EH0145-3-PWR, pp. 36-46 (1979).

D. L. Hillhouse, 0. Petersons, and W. C. Sze, "A Prototype System for On-Site

Calibration of Coupling Capacitor Voltage Transformers (CCVTs)," IEEE Trans.

Power Appar. Syst. PAS-98 , No. 3, pp. 1026-1036 (May/June 1979).

R. L. Kahler, "An Electronic Ratio Error Set for Current Transformer

Calibrations," IEEE Trans. Instrum. Meas. IM-28 , No. 2, pp. 162-164

(June 1979).

F. R. Kotter, "Characterization of the Electric Environment Under HVDC

Transmission Lines: Instrumentation and Measurement Techniques," Proceedings

of Workshop on Electrical and Biological Effects Related to HVDC Transmission,

Richland, Washington, October 19-20, 1978 (Aug. 1979).

R. E. Hebner and S. Annestrand, "Evaluation of Calibration Techniques for

Multimegavolt Impulse Dividers," International Symposium on High Voltage

Engineering, Milan, Italy, pp. 1-4 (Aug. 1979).
R. E. Hebner, D. L. Hillhouse, and R. A. Bullock, "Evaluation of a Multimegavolt Impulse Measurement System," NBSIR 79-1933, 102 pages

(Nov. 1979).

M. Misakian, "Generation and Measurement of DC Electric Fields With Space

Charge," NBSIR 80-2177, 39 pages (Jan. 1981).

D. L. Hillhouse, "Guide for Safe Operating Procedures at High Voltage

Substations by NBS and Utility Staff During the Field Calibration of Coupling

Capacitor Voltage Transformers (CCVTs)," NBSIR 81-2192, 10 pages (Feb. 1981).

[Limited distribution; not available from NTIS]

W. E. Anderson and J. D. Ramboz, "1980 Annual Report: Technical Contributions

to the Development of Incipient Fault Detection/Location Instrumentation,"

NBSIR 81-2235, 125 pages (March 1981).

R. E. Hebner, "Development of Power System Measurements—Quarterly Report

January 1, 1981 to March 31, 1981," NBSIR 81-2283, 14 pages (Apr. 1981).

R. H. McKnight, F. R. Kotter, M. Misakian, and P. Ortiz, "1980 Annual Report:

Electric and Magnetic Field Measurements," NBSIR 81-2267, 65 pages (May 1981).

M. Misakian, "Generation and Measurement of DC Electric Fields With Space

Charge," J. Appl . Phys. 52, No. 5, pp. 3135-3144 (May 1981).

R. E. Hebner, "Development of Power System Measurements—Quarterly Report

April 1, 1981 to June 30, 1981," NBSIR 81-2334, 15 pages (June 1981).

D. A. Leep and F. R. Kotter, "Discussion of Paper No. 81 WM 126-2, Measuring

Voltage and Current Harmonics on Distribution Systems," by M. F. McGranaghan,

J. H. Shaw, and R. E. Owen, IEEE Trans. Power Appar. System. PAS- 100 , No. 7,

pp. 3599-3608 (July 1981).

R. J. Van Brunt, "Discussion of paper by J. M. Pelletier et al.," IEEE Trans.

Power Appar. Syst. PAS- 100 , No. 8, pp. 3867-3868 (Aug. 1981).

F. R. Kotter and M. Misakian, "AC Transmission Line Field Measurements,"

NBSIR 77-1311, 79 pages (Oct. 1981).

D. L. Hillhouse and D. A. Leep, "Analysis of the Calibration of Metering CCVTs

in a Utility Substation," NBSIR 81-2360, 53 pages (Oct. 1981).

N. M. Oldham and R. S. Turgel , "A Power Factor Standard Using Digital Waveform

Generation," IEEE Trans. Power Appar. Syst. PAS-100 , No. 11, pp. 4435-4438

(Nov. 1981).

R. H. McKnight, F. R. Kotter, and M. Misakian, "Measurement of Ion Current

Density at Ground Level in the Vicinity of High Voltage DC Transmission

Lines," NBSIR 81-2410, 27 pages (Dec. 1981).

R. E. Hebner, "Development of Power System Measurements— Quarterly Report

October 1, 1981 to December 31, 1981," NBSIR 82-2501, 21 pages (May 1982).

712
R. E. Hebner, "Experimental Comparison of Step-Response and Ramp-Response

Measurements in Freestanding Dividers, Measurement of Electrical Quantities in

Pulse Power Systems," Nat. Bur. Stand. (U.S.), Spec. Publ. 628, pp. 26-33

(June 1982). [GPO]

"Measurement of Electrical Quantities in Pulse Power Systems," Proceedings of

the Workshop on Measurement of Electrical Quantities in Pulse Power Systems,

Boulder, CO, March 2-4, 1981, R. H. McKnight and R. E. Hebner, Jr., editors,

Nat. Bur. Stand. (U.S.), Spec. Publ. 628, 410 pages (June 1982). [GPO]

R. E. Hebner, "Development of Power System Measurements—Quarterly Report

January 1, 1982 to March 31, 1982," NBSIR 82-2528, 20 pages (June 1982).

R. H. McKnight and F. R. Kotter, "A Facility to Produce Uniform Space Charge

for Evaluating Ion Measuring Instruments," NBSIR 82-2517, 33 pages

(June 1982).

R. E. Hebner, "Development of Power System Measurements—Quarterly Report

April 1, 1982 to June 30, 1982," NBSIR 82-2586, 20 pages (Oct. 1982).

R. H. McKnight, F. R. Kotter, and M. Misakian, "Measurement of Ion Current

Density at Ground Level in the Vicinity of High Voltage DC Transmission

Lines," IEEE Trans. Power Appar. Syst. PAS-102 , No. 4, pp. 934-941

(April 1983).

R. E. Hebner, "Development of Power System Measurements—Quarterly Report

July 1, 1982 to September 30, 1982," NBSIR 83-2705, 25 pages (May 1983).

R. H. McKnight and F. R. Kotter, "A Facility to Produce Uniform Space Charge

for Evaluating Ion Measuring Instruments," IEEE Trans. Power Appar. Syst.,

PAS-102 , No. 7, pp. 2349-2357 (July 1983).

E. F. Kelley and R. E. Hebner, "Electro-Optic Measurement of the Electric

Field Distribution in Transformer Oil," IEEE Trans. Power Appar. Syst.,

PAS-102 , No. 7, pp. 2092-2097 (July 1983).

J. D. Ramboz and R. C. McAuliff, "A Calibration Service for Wattmeters and

Watthour Meters," Nat. Bur. Stand. (U.S.), Tech. Note 1179, 111 pages

(July 1983). [GPO]

R. H. McKnight and H. K. Schoenwetter, "Evaluation of Transient Measurement

Methods in Gas-Insulated Transmission Lines," NBSIR 83-2753, 75 pages

(August 1983).

R. E. Hebner, "Development of Power System Measurements— Quarterly Report

October 1, 1982 to December 31, 1982," NBSIR 83-2755, 41 pages

(Sept. 1983).

R. E. Hebner, "Development of Power System Measurements— Quarterly Report

January 1, 1983 to March 31, 1983," NBSIR 83-2761, 27 pages (Oct. 1983).

M. Misakian and P. M. Fulcomer, "Measurement of Nonuniform Power Frequency

Electric Fields," IEEE Trans, on Electr. Insul. EI-18 , No. 6, pp. 657-661,

December 1983.

713
High Voltage Measurements

E. C. Cassidy and S. Abramowitz, "Time-Resolved Emission and Absorption

Studies of Exploding Wire Spectra," Proc. 4th Conf. Exploding Wires,

Book, Exploding Wires 4, pp. 109-124 (Plenum Press, New York, NY, 1968).

E. C. Cassidy, "Time-Resolved Studies of Spectra Produced by Electrically

Exploded Wires," Die Naturwissenschaften, Heft 3, pp. 125-128 (1968).

H. E. Radford and K. M. Evenson, "Paramagnetic-Resonance Spectrum of

Metastable (?D) Atomic Nitrogen," Phys. Rev. J68, No. 1, pp. 70-74

(Apr. 5, 1968).

0. Petersons, "A Transformer-Ratio-Arm Bridge for Measuring Large

Capacitors Above 100 Volts," IEEE Trans. Power Appar. Syst. PAS -87 , No. 5, pp. 1354-1361 (May 1968).

E. C. Cassidy and H. N. Cones, "Calibration of a Kerr Cell System

for High Voltage Pulse Measurements," Final Report for Sandia

Laboratories, Sandia Report No. SC-CR-68-3730 (Aug. 1968).

E. C. Cassidy, S. Abramowitz, and C. W. Beckett, "Investigations of

the Exploding Wire Process as a Source for High Temperature Studies,"

Nat. Bur. Stand. (U.S.), Monogr. 109, 53 pages (Nov. 1968).

H. E. Radford, "Scanning Microwave Echo Box Spectrometer," Rev. Sci

Instrum. 39, No. 11, pp. 1687-1691 (Nov. 1968).

E. C. Cassidy, H. N. Cones, 0. C. Wunsch, and S. R. Booker, "Calibration

of a Kerr Cell System for High-Voltage Pulse Measurements," IEEE Trans.

Instrum. Meas. IM-17 , No. 4, pp. 313-320 (Dec. 1968).

E. C. Cassidy and H. N. Cones, "Development and Analysis of Techniques

for Calibration of Kerr Cell Pulse-Voltage Measuring Systems III,"

Final Report for Sandia Laboratories, NBS Report No. 10 078 (Aug. 1969).

E. C. Cassidy and H. N. Cones, "Development and Analysis of Techniques

for Calibration of Kerr Cell Pulse-Voltage Measuring Systems IV,"

Final Report for Sandia Laboratories, NBS Report No. 10 296 (Aug. 6, 1970).

E. C. Cassidy, H. N. Cones, and S. R. Booker, "Development and Evaluation

of Electrooptical High-Voltage Pulse Measurement Techniques," IEEE Trans.

Instrum. Meas. IM-19 , No. 4, pp. 395-402 (Nov. 1970).

H. E. Radford and C. V. Kurtz, "Stark Effect and Hyperfine Structure

of HCN Measured With an Electric Resonance" Maser Spectrometer," J. Res.,

Nat. Rur. Stand. (U.S.), (Phys. and Chem.) 74A , No. 6, pp. 791-799

(Nov. -Dec. 1970).

714
E. C. Cassidy, "Development and Analysis of Techniques for Calibration

of Kerr Cell Pulse-Voltage Measuring Systems V," Final Report for Sandia

Laboratories, NBS Report No. 10 493 (Oct. 1972).

D. L. Hillhouse, "Circuit for Impulse Testing of Gas-Tube Lightning

Arresters," IEEE Trans. Commun. COM-20 , No. 5, pp. 936-941 (Oct. 1 972).

E. C. Cassidy, R. E. Hebner, W. E. Anderson, R. J. Sojka, and S. R. Booker,

"Development and Analysis of Techniques for Calibration of Kerr Cell PulseVoltage Measuring Systems VI," Final Report for Sandia Laboratories,

NBS Report No. 10 945 (Nov. 1, 1972).

A. Erez, "Low-Frequency Electrical Signal Measurement by Electroopti cal

Methods," IEEE Trans. Instrum. Meas. IM-21 , No. 4, pp. 358-360 (Nov. 1972).

E. C. Cassidy, W. E. Anderson, and S. R. Booker, "Recent Refinements and

Developments in Kerr System Electrical Measurement Techniques," IEEE

Trans. Instrum. Meas. IM-21 , No. 4, pp. 504-510 (Nov. 1972).

R. E. Hebner, Jr. and E. C. Cassidy, "Measurement of 60 Hz Voltages Using

the Kerr Effect," Rev. Sci . Instrum. 43, No. 12, pp. 1839-1841 (Dec. 1972).

E. C. Cassidy, R. E. Hebner, R. J. Sojka, and M. Zahn, "Development and

Analysis of Techniques for Calibration of Kerr Cell Pulse-Voltage Measuring

Systems VII," Final Report for Sandia Laboratories, N8SIR 73-4U3, 130 pages

(Nov. 1973) (no copies available).

D. L. Hillhouse and A. E. Peterson, "A 300-kV Compressed Gas Standard

Capacitor With Negligible Voltage Dependence," IEEE Trans. Instrum.

Meas. IM-22 , No. 4, pp. 408-416 (Dec. 1973).

R. E. Hebner, E. C. Cassidy, and R. J. Sojka, "Development and Analysis

of Techniques for Calibration of Kerr Cell Pulse-Voltage Measuring

Systems VIII," NBSIR 74-564, 55 pages (Aug. 21, 1974).

R. E. Hebner, Jr., "Calibration of Kerr Systems Used to Measure High Voltage

Pulses," NBSIR 75-774, 51 pages (Aug. 7, 1975).

R. E. Hebner, Jr., "Electrical Measurement of High Voltage Pulses in Diagnostic

X-Ray Units/' NBSIR 75-775, 62 pages (Nov. 1 , 1975).

R. E. Hebner, Jr. and S. R. Booker, "A Portable Kerr System for the Measurement

of High Voltage Pulses," Proc. IEEE SOUTHEAST CON 75, PP- 3A-1-1 - 3A-5-1

(1975).

0. Petersons and W. E. Anderson, "A Wide-Range High-Voltage Capacitance

Bridge With One PPM Accuracy," IEEE Trans. Instrum. Meas. IM-24 , No. 4,

pp. 336-344 (Dec. 1975).

715
R. E. Hebner, E. C. Cassidy, and J. E. Jones, "Improved Techniques for the

Measurement of High-Voltage Impulses Using the Electrooptic Kerr Effect,"

IEEE Trans. Instrum. Meas. IM-24 , No. 4, pp. 361-366 (Dec. 1975).

R. L. Kahler, D. L. Hillhouse, and W. E. Anderson, "A Simple Overcurrent

Protection Circuit for a High Voltage Laboratory," IEEE Trans. Instrum.

Meas., pp. 161-162 (June 1976).

C. B. Childers, R. F. Dziuba, and L. H. Lee, "A Resistive Ratio Standard

for Measuring Direct Voltages to 10 kV," IEEE Trans. Instrum. Meas. IM-25 , No. 4, pp. 505-508 (Dec. 1976).

R. E. Hebner, Jr. and M. Misakian, "Calibration of High-Voltage Pulse

Measurement Systems Based on the Kerr Effect," NBSIR 77-1317, 38 pages

(Sept. 1 6, 1977).

R. E. Hebner, Jr., R. A. Malewski, and E. C. Cassidy, "Optical Methods of

Electrical Measurement at High Voltage Levels," Proc. IEEE, J55_, No. 11,

pp. 1524-1548 (Nov. 1977).

D. L. Hillhouse and W. C. Sze, "Calibration of CCVTs in the Substation,"

45th Annual International Conference of Doble Clients, Section 9-501 (1978).

M. Misakian, F. R. Kotter, and R. L. Kahler, "Miniature ELF Electric

Field Probe," Rev. Sci. Instrum. 49, No. 7, pp. 933-935 (July 1978).

W. E. Anderson, R. S. Davis, 0. Petersons, and W.J.M. Moore, "An

International Comparison of High Voltage Capacitor Calibrations," IEEE

Trans. Power Appar. Syst. PAS-97 , No. 4, pp. 1217-1223 (July /Aug. 1978).

D. L. Hillhouse, 0. Petersons, and W. C. Sze, "A Prototype System for

On-Site Calibration of Coupling Capacitor Voltage Transformers (CCVTs),"

IEEE Trans. Power Appar. Syst. PAS -98 , No. 3, pp. 1026-1036 (May/June 1979).

R. E. Hebner and S. Annestrand, "Evaluation of Calibration Techniques

for Multimegavolt Impulse Dividers," International Symposium on High

Voltage Engineering, Milan, Italy, pp. 1-4 (Aug. 1979).

R. E. Hebner and M. Misakian, "Temperature Dependence of the Electrooptic Kerr Coefficient of Nitrobenzene," J. Appl . Phys. 50(9) , pp. 6016-6017

(Sept. 1979).

R. E. Hebner, D. L. Hillhouse, and R. A. Bullock, "Evaluation of a Multimegavolt Impulse Measurement System," NBSIR 79-1933, 102 pages

(Nov. 1979).

R. H. McKnight and R. E. Hebner, "X-CAL -- A Calibration System for

Electrical Measurement Devices Used With Diagnostic X-Ray Units,"

NBSIR 80-2072, 75 pages (June 1980).

716
D. L. Hillhouse, "Guide for Safe Operating Procedures at High Voltage

Substations by NBS and Utility Staff During the Field Calibration of

Coupling Capacitor Voltage Transformers (CCVTs), NBSIR 81-2192, 10 pages

(Feb. 1981). [Limited distribution; not available from NTIS]

D. L. Hillhouse and D. A. Leep, "Analysis of the Calibration of Metering

CCVTs in a Utility Substation," NBSIR 81-2360, 53 pages (Oct. 1981).

D. L. Hillhouse, 0. Petersons, and W. C. Sze, "A Simplified System for

Calibration of CCVTs in the Substation," Nat. Bur. Stand. (U.S.), Tech.

Note 1155, 57 pages (May 1982).

J. 0. Ramboz, A. R. Ondrejka, and W. E. Anderson, "Sampling-Rate Drift

Problems in Transfer Function Analysis of Electrical Power Cables,"

Proceedings of the Waveform Recorder Seminar, Boulder, CO, October 1981,

Nat. Bur. Stand. (U.S.), Spec. Publ. 634, pp. 47-53 (June 1982). [GPO]

"Measurement of Electrical Quantities in Pulse Power Systems," Proceedings

of the Workshop on Measurement of Electrical Quantities in Pulse Power Systems,

Boulder, CO, March 2-4, 1981, R. H. McKnight and R. E. Hebner, Jr., editors,

Nat. Bur. Stand. (U.S.), Spec. Publ. 628, 410 pages (June 1982). [GPO]

W. E.. Anderson, J. D. Ramboz, and A. R. Ondrejka, "The Detection of

Incipient Faults in Transmission Cables Using Time Domain Ref lectometry

Techniques: Technical Challenges," IEEE Trans. Power Appar. Syst.

PAS-101 , No. 7, pp. 1928-1934 (July 1982).

D. L. Hillhouse, "Effects of High-Voltaqe Switching on the EPRI-NBS

Coupling Capacitor Voltage Transformer (CCVT) Calibration System Standard

Divider," NBSIR 83-2666, 37 pages (March 1983).

R. H. McKnight and H. K. Schoenwetter, "Evaluation of Transient Measurement

Methods in Gas-Insulated Transmission Lines," NBSIR 83-2753, 75 pages

(August 1983).

i

717
Data Acquisition and Conversion

S. K. Tewksbury, F. C. Meyer, D. C. Rollenhagen, H. K. Schoenwetter,

and T. M. Souders, "Terminology Related to the Performance of S/H, A/D,

and D/A Circuits," IEEE Trans. Circuits Syst. CAS-25 , No. 7, pp. 419-426

(July 1978).

B. F. Field, "A Fast Response Low-Frequency Voltmeter," IEEE Trans.

Instrum. Meas. IM-27 , No. 4, pp. 368-372 (Dec. 1978).

T. M. Souders, "A Bridge Circuit for the Dynamic Characterization of

Sample/Hold Amplifiers," IEEE Trans. Instrum. Meas. IM-27 , No. 4,

pp. 409-413 (Dec. 1978).

H. K. Schoenwetter, "A High-Speed Low-Noise 18-Bit Di gi tal -to-Analog

Converter," IEEE Trans. Instrum. Meas. IM-27 , No. 4, pp. 413-417

(Dec. 1978).

T. Mo Souders and D. R. Flach, "A 20 Bit + Sign, Relay Switched D/A

Converter," Nat. Bur. Stand. (U.S.), Tech. Note 1105, 21 pages

(Oct. 1979). [GPO]

T. M. Souders and D. R. Flach, "An Automated Test Set for High Resolution

Analog-to-Di gital and Di gi tal -to-Analog Converters," IEEE Trans. Instrum.

Meas. IM-28 , No. 4, pp. 239-244 (Dec. 1979).

A. G. Perrey and H. K. Schoenwetter, "A Schottky Diode Bridge Sampliny

Gate," Nat. Bur. Stand. (U.S.), Tech. Note 1121, 18 pages (May 1980).

[GPO]

T. M. Souders and J. A. Lechner, "A Technique for Measuring the Equivalent

RMS Input Noise of A/D Converters," IEEE Trans. Instrum. Meas. IM-29 , No. 4,

pp. 251-256 (Dec. 1980).

T. M. Souders, D. R. Flach, and B. A. Bell, "A Calibration Service for

Analog-to-Di gital and Di gital -to-Analog Converters," Nat. Bur. Stand. (U.S.),

Tech. Note 1145, 66 pages (July 1981). [GPO]

T. M. Souders and D. R. Flach, "An NBS Calibration Service for A/D and D/A

Converters," 1981 International Test Conference Digest, pp. 290-303 (Oct. 1981).

T. M. Souders, "A Dynamic Test Method for High-Resolution A/D Converters,"

IEEE Trans. Instrum. Meas. IM-31 , No. 1, pp. 3-5 (March 1982).

718
T. M. Souders and D. R. Flach, "Measurement of the Transient Versus

Steady-State Response of Waveform Recorders," Proceedings of the Waveform

Recorder Seminar, Boulder, CO, October 1981, Nat. Bur. Stand. (U.S.), Spec.

Publ. 634, pp. 27-34 (June 1982). [GPO]

D. R. Flach, "Steady-State Tests of Waveform Recorders," Proceedings of the

Waveform Recorder Seminar, Boulder, CO, October 1981, Nat. Bur. Stand. (U.S.),

Spec. Publ. 634, pp. 7-21 (June 1982). [GPO]

J. R. Andrews, N. S. Nahman, and B. A. Bell, "Status of Reference Waveform

Standards Development at NBS," Proceedings of the Waveform Recorder Seminar,

Boulder, CO, October 1981, Nat. Bur. Stand. (U.S.) , Spec. Publ. 634, pp. 69-88

(June 1982). [GPO]

J. D. Ramboz, A. R. Ondrejka, and W. E. Anderson, "Sampling-Rate Drift

Problems in Transfer Function Analysis of Electrical Power Cables,"

Proceedings of the Waveform Recorder Seminar, Boulder, CO, October 1981, Nat.

Bur. Stand. (U.S.), Spec. Publ. 634, pp. 47-53 (June 1982). [GPO]

B. A. Bell, B. F. Field, and T. H. Kibalo, "A Fast Response, Low-Frequency

Sampling Voltmeter," Nat. Bur. Stand. (U.S.), Tech. Note 1159, 113 pages

(Aug. 1982). [GPO]

T. M. Souders, D. R. Flach, and T. C. Wong, "An Automated Test Set for the

Dynamic Characterization of A/D Converters," IEEE Trans. Instrum. Meas. IM-32 , No. 1, pp. 180-186 (March 1983).

H. K. Schoenwetter, "High Accuracy Settling Time Measurements," IEEE Trans.

Instrum. Meas. IM-32 , No. 1, pp. 22-27 (March 1983).

B. A. Bell and A. G. Perrey, "Peak Conductance Measurements of GaAs

Switching Devices," Picosecond Optoelectronics, Gerard Mourou, ed.,

Proc. SPIE 439, pp. 128-139 (Aug. 1983).

J. R. Andrews, B. A. Bell, and E. E. Baldwin, "Reference Flat Pulse

Generator," Nat. Bur. Stand. (U.S.), Tech. Note 1067, 72 pages

(Oct. 1983). [GPO]

H. K. Schoenwetter, "A Programmable Voltage-Step Generator for Testing

Waveform Recorders," IEEE Instrum. and Meas. Technology Conference

Proceedings, pp. 71-72 (Jan. 1984).

719
Automatic Testing

B. A. Bell, T. M. Souders, B. C. Belanger, and R. A. Kamper, "Challenges in

Achieving ATE Traceability to NBS," Proc. IEEE AUTOTESTCON , pp. 233-238

(Sept. 1979).

B. A. Bell, "Precision Electronic Test Equipment Calibration Standards at

NBS," Proc. ATE Seminar and Test Instruments Conference, pp. 138-168

(Jan. 1980).

K. J. Lentner, C. B. Childers, and S. G. Tremaine, "A Semiautomatic System for

AC/DC Difference Calibration," IEEE Trans. Instrum. Meas. IM-29 , No. 4,

pp. 400-405 (Dec. 1980).

M. G. Buehler and L. W. Linholm, "Toward a Standard Test Chip Methodology for

Reliable, Custom Integrated Circuits," Proc. of the Custom Integrated Circuit

Conf., pp. 142-146 (May 1981).

M. G. Buehler and D. S. Perloff, "Microelectronic Test Chip and Associated

Parametric Testers: Present and Future," Semiconductor Silicon '81,

H. R. Huff, R. J. Kriegler, and Y. Takeishi, eds., Electrochemical Society,

pp. 859-867 (May 1981).

B. A. Bell and 0. Petersons, "ATE Calibration by Means of Dynamic Transport

Standards," IEEE AUT0TESTC0N '81 Proceedings, pp. 280-287 (Oct. 1981).

K. J. Lentner and S. G. Tremaine, "A Semiautomatic AC/DC Thermal Voltage

Converter Calibration System," NBSIR 82-2576, 62 pages (Sept. 1982).

W. G. Eicke, Jr., T. F. Leedy, B. R. Moore, and C. F. Brown, "Measurement

Assurance Programs in a Field Environment," Proc. National Conference of

Standards Laboratories, 1982 Workshop and Symposium, Gaithersburg, Md.,

Oct. 4-7, 1982, pp. E-l-E-9.

720
Electrical Insulation Measurements

H. E. Radford, "Electrical Breakdown in Ammonia at Low Pressure,"

NBS Report No. 9882, 12 pages (July 22, 1968).

E. C. Cassidy and H. N. Cones, "A Kerr Electro-Optical Technique for

Observation and Analysis of High-Intensity Electric Fields," J. Res.,

Nat. Bur. Stand. (U.S.), (Eng. & Instr.) 73C, Nos. 1 and 2, pp. 5-13

(Jan. -June 1969).

E. C. Cassidy and H. N. Cones, "Use of Expanded Laser Beam to Analyze

High-Intensity Electric Fields," J. Soc. Motion Pict. Telev. Eng. 19,

No. 7, pp. 590-591 (July 1970).

E. C. Cassidy and H. N. Cones, "Electro-Optical Observations and

Measurements of Distorted High-Intensity Electric Fields," Proc.

1969 Conf. on Electrical Insulation and Dielectric Phenomena

(Buck Hill Falls, Pa., Oct. 20-22, 1969), pp. 77-86 (National Academy

of Sciences, Washington, DC, 1970).

E. C. Cassidy, "Pulsed Laser Kerr System Polarimeter for Electro-Optical

Fringe Pattern Measurement of Transient Electrical Parameters," Rev.

Sci. Instrum. 43, No. 6, pp. 886-893 (June 1972).

E. C. Cassidy and R. E. Hebner, "Experimental Study of the Behavior

of Nitrobenzene Under Varied High Voltage Conditions," 1972 Annual

Report, Conference on Electrical Insulation and Dielectric Phenomena,

pp. 37-44 (1973).

R. E. Hebner, Jr., E. C. Cassidy, M. Zahn, and R. J. Sojka, "Electric

Field Distributions and Space Charge Behavior in Nitrobenzene Under Low

Frequency Alternating Voltage, 1973 Annual Report," Conference on Electrical Insulation and Dielectric Phenomena, pp. 112-119 (1974).

E. C. Cassidy, R. E. Hebner, Jr., M. Zahn, and R. J. Sojka, "Kerr-Effect

Studies of an Insulating Liquid Under Varied High-Voltage Conditions,"

IEEE Trans. Electr. Insul. EI-9 , No. 2, pp. 43-56 (June 1974).

R. E. Hebner, Jr., R. J. Sojka, and E. C. Cassidy, "Kerr Coefficients of

Nitrobenzene and Water," NBSIR 74-544, 35 pages (Aug. 7, 1974).

F. R. Kotter, "A Study of Air-Gap Breakdown at 28.5 Kilohertz,"

NBSIR 75-731, 42 pages (June 20, 1975).

D. B. Miller (Principal Investigator), V. E. Bower, F. R. Kotter,

0. Petersons, M. M. Birky, C. M. Huggett, and A. J. Macek, "An Appraisal

of Tests and Standards for the Evaluation of Electrical Insulating Fluids,

NBSIR 76-1054, 124 pages (May 1976).

721
M. Misakian and R. E. Hebner, Jr., "Kerr Coefficients of Polychlori nated

Biphenyls and Chlorinated Naphthalene," J. Appl . Phys. 47, No. 9,

pp. 40S2-4055 (Sept. 1976).

W. E. Anderson, R. S. Davis, F. I. Mopsik, S. J. Kryder, F. Khoury,

J. P. Colson, and L. H. Bolz, "Measurements on Insulating Materials

at Cryogenic Temperatures," ERDA Report No. CONS/2062-1 (Sept. 1976).

W. E. Anderson and R. S. Davis, "Measurement of AC Insulation Losses

at Cryogenic Temperatures," IEEE Trans. Electr. Insul. EI-12 , No. 1,

pp. 51-54 (Feb. 1977).

R. E. Hebner, Jr., R. A. Malewski, and E. C. Cassidy, "Optical Methods

of Electrical Measurement at High Voltage Levels," Proc. IEEE, 65_, No. 11,

pp. 1524-1548 (Nov. 1977).

E, F. Kelley and R. E. Hebner, "Measurement of Prebreakdown Electric

Fields in Liquid Insulants," 1978 Annual Report, Conference on Electrical

Insulation and Dielectric Phenomena, National Academy of Sciences,

Washington, DC, pp. 206-212 (Oct. 1978).

E. F. Kelley and R. E. Hebner, "Time Evolution of the Electric Field

Associated With Breakdown Phenomena in Liquids," 1979 Annual Report,

Conference on Electrical Insulation and Dielectric Phenomena, pp. 203-211

(1979) .

R. J. Van Brunt, "Minutes of Workshop on Gaseous Dielectrics for Use in

Future Electric-Power Systems - September 10-11, 1979," NBSIR 80-1966

(Jan. 1980). [Limited distribution; not available from NTIS]

W. E. Anderson and R. S. Davis, "Measurements on Insulating Materials

at Cryogenic Temperatures," NBSIR 79-1950, 164 pages (Jan. 1980).

R. J. Van Brunt, J. S. Hilten, and D. P. Silver, "Partial-Discharge

Pulse Height Distributions and Frequencies for Positive and Negative

DC Corona in SF5 and SF6-N2 Mixtures," Proc. 2nd Intl. Symp. on Gaseous Dielectrics, Gaseous Dielectrics II , Ed. by L. G. Christophorou,

Pergamon Press, NY, pp. 303-311 (March 1980).

D. A. Leep and R. J. Van Brunt, "Safety Considerations for Handling

SF 6 Used in Electrical Equipment," NBSIR 80-2063 (June 1980).

[Limited distribution; not available from NTIS]

E. F. Kelley and R. E. Hebner, "Breakdown Between Bare Electrodes

With an Oil-Paper Interface," NBSIR 80-2071, 33 pages (June 1980).

R. J. Van Brunt and M. Misakian, "Comparison of DC and 60 Hz AC Positive

and Negative Partial Discharge Inceptions in SF6, 1980 Annual Report,"

Conference on Electrical Insulation and Dielectric Phenomena, pp. 461-469

(1980) .

722
E. F. Kelley and R. E. Hebner, "The Electric Field Distribution Associated

With Prebreakdown Phenomena in Nitrobenzene," J. Appl. Phys. b»2, No. 1,

pp. 191-195 (Jan. 1931).

M. Misakian, "Generation and Measurement of DC Electric Fields With

Space Charge," NBSIR 80-2177, 39 pages (Jan. 1981).

E. F. Kelley and R. E. Hebner, "Prebreakdown Phenomena Between Sphere-Sphere

Electrodes in Transformer Oil," Appl. Phys. Lett. 38, No. 4, pp. 231-233

(Feb. 1981).

W. E. Anderson and J. D. Ramboz, "1980 Annual Report: Technical

Contributions to the Development of Incipient Fault Detection/Location

Instrumentation," NBSIR 81-2235, 125 pages (March 1981).

R. J. Van Brunt, M. Misakian, D. A. Leep, K. J. Moy, and E. C. Beaty,

"1980 Annual Report: Technical Assistance for Future Insulation Systems

Research," NBSIR 81-2242, 96 pages (Apr. 1981).

R. E. Hebner, E. F. Kelley, J. E. Thompson, T. S. Sudershan, and T. B. Jones,

"1980 Annual Report: Optical Measurements for Interfacial Conduction and

Breakdown," NBSIR 81-2275, 83 pages (May 1981).

R. H. McKnight, F. R. Kotter, M. Misakian, and P. Ortiz, "1980 Annual

Report: Electric and Magnetic Field Measurements," NBSIR 81-2267, 65 pages

(May 1981).

M. Misakian, "Generation and Measurement of DC Electric Fields With Space

Charge," J. Appl. Phys. 52, No. 5, pp. 3135-3144 (May 1981).

R. E. Hebner, E. F. Kelley, E. 0. Forster, and G. J. FitzPatrick,

"Observation of Prebreakdown and Breakdown Phenomena 1n Liquid Hydrocarbons,"

7th Inter. Conf. on Conduction and Breakdown in Dielectric Liquids, Berlin,

West Germany, pp. 177-181 (July 1981).

M. Zahn, E. 0. Forster, E. F. Kelley, and R. E. Hebner, "Hydrodynamic Shock

Wave Propagation After Electrical Breakdown," 7th Inter. Conf. on Conduction

and Breakdown in Dielectric Liquids, Berlin, West Germany, pp. 398-403

(July 1981).

D. A. Leep and F. R. Kotter, "Discussion of Paper No. 81 WM 126-2,

Measuring Voltage and Current Harmonics on Distribution Systems, by

M. F. McGranaghan, J. H. Shaw, and R. E. Owen," IEEE Trans, on Power

Appar. Syst. PAS -100 , No. 7, pp. 3599-3608 (July 1981).

723
E. F. Kelley and R. E. Hebner, "Electrical Breakdown in Composite Insulating

Systems: Liquid-Solid Interface Parallel to the Field," IEEE Trans, on Electrical Insulation EI- 16 , No. 4, pp. 297-303 (Aug. 1981).

R. J. Van Brunt, "Discussion of paper by J. M. Pelletier et al.," IEEE Trans.

Power Appar. Syst. PAS-100 , pp. 3867-3868 (Aug. 1981).

R. E. Hebner, E. F. Kelley, E. 0. Forster, and G. J. FitzPatrick,

"Observations of Prebreakdown and Breakdown Phenomena in Liquid Hydrocarbons

II. Point-Plane Geometry, 1981 Annual Report," Conference on Electrical

Insulation and Dielectric Phenomena, pp. 377-389 (Oct. 1981).

F. R. Kotter and M. Misakian, "AC Transmission Line Field Measurements,"

NBSIR 77-1311, 79 pages (Oct. 1981).

R. J. Van Brunt and D. A. Leep, "Characterization of Point-Plane Corona Pulses

in SF 6

," J. Appl. Phys. 52, No. 11, pp. 6588-6600 (Nov. 1981).

R. H. McKnight, F. R. Kotter, and M. Misakian, "Measurement of Ion Current

Density at Ground Level in the Vicinity of High Voltage DC Transmission

Lines," NBSIR 81-2410, 27 pages (Dec. 1981).

R. J. Van Brunt, "Discussion on 81 SM 322-7 Breakdown of Rod-Plane Gaps in

SF5 Under Positive Switching Impulses, by H. Anis and K. Srivastava," IEEE

Trans. Power Appar. Syst. PAS- 101 , No. 3, p. 546 (March 1982).

R. J. Van Brunt and M. Misakian, "Mechanisms for Inception of DC and 60-Hz AC

Corona in SFs," IEEE Trans. Electr. Insul. EI-17 , No. 2, pp. 106-120

(April 1982).

R. H. McKnight, "The Measurement of Net Space Charge Density Using Air

Filtration Methods," NBSIR 82-2486, 28 pages (April 1982).

'

E. F. Kelley, R. E. Hebner, E. 0. Forster, and G. J. FitzPatrick,

"Observations of Pre- and Post-Breakdown Events in Polydimethylsiloxanes,"

Proceedings of the 1982 IEEE International Symposium on Electrical Insulation,

No. 82CH1780-6-E1, pp. 255-258 (June 1982).

'

R. H. McKnight, F. R. Kotter, M. Misakian, and J. N. Hagler, "1981 Annual

Report— Electrical and Magnetic Field Measurements," NBSIR 82-2527, 48 pages

(July 1982).

R. E. Hebner, E. F. Kelley, E. 0. Forster, and G. J. FitzPatrick, "Observation

of Prebreakdown and Breakdown Phenomena in Liquid Hydrocarbons," Journal of

Electrostatics, Vol. 12, pp. 265-283, 1982.

M. Zahn, E. 0. Forster, E. F. Kelley, and R. E. Hebner, "Hydrodynamic Shock

Wave Propagation After Electrical Breakdown," Journal of Electrostatics,

Vol. 12, pp. 535-546, 1982.

R. J. Van Brunt and D. A. Leep, "Corona-Induced Decomposition of SF6,"

Proceedings of the Third International Symposium on Gaseous Dielectrics,

Gaseous Dielectrics III, pp. 402-409, 1982.

724
W. F. Schmidt and R. J. Van Brunt, "Comments on the Effect of Electron

Detachment in Initiating Breakdown in Gaseous Dielectrics," Proceedings of the

Third International Symposium on Gaseous Dielectrics, Gaseous Dielectrics III,

pp. 561-563, 1982.

R. J. Van Brunt, "Effects of H2O on the Behavior of SFs Corona,"

Proceedings of the Seventh International Conference on Gas Discharges and

Their Applications, London, England, 1982, pp. 255-258.

G. J. FitzPatrick, E. 0. Forster, E. F. Kelley, and R. E. Hebner, "Effects of

Chemical Impurities on Prebreakdown Events in Toluene, 1982 Annual Report,"

Conference on Electrical Insulation and Dielectric Phenomena, Oct. 17-21,

1982, 464-472, October 1982.

R. J. Van Brunt, M. Misakian, D. A. Leep, E. C. Beaty, J. W. Gallagher,

C. M. Cooke, K. Wyatt, and R. G. Gels, "1981 Annual Report—Technical

Assistance for Future Insulation Systems Research," NBSIR 82-2555, 157 pages

(Nov. 1982).

R. E. Hebner, E. F. Kelley, and J. N. Hagler, "1981 Annual Report: Optical

Measurements for Interfacial Conduction and Breakdown," NBSIR 82-2629,

82 pages (Jan. 1983).

R. J. Van Brunt and M. Misakian, "Role of Photodetachment in Initiation of

Electric Discharges in and O2," J. Appl . Phys., Vol. 54, No. 6,

pp. 3074-3079, June 1983.

R. H. McKnight and F. R. Kotter, "A Facility to Produce Uniform Space Charge

for Evaluating Ion Measuring Instruments," IEEE Trans. Power Appar. Syst.,

PAS-102 , No. 7, pp. 2349-2357 (July 1983).

E. F. Kelley and R. E. Hebner, Electro-Optic Measurement of the Electric Field

Distribution in Transformer Oil," IEEE Trans. Power Appar. Syst. PAS-102 , No. 7, pp. 2092-2097 (July 1983).

R. E. Hebner, E. F. Kelley, G. J. FitzPatrick, and E. 0. Forster, "The Effect

of Impurities on Positive Streamer Propagation in n-Hexane," 1983 Annual

Report, Conference on Electrical Insulation and Dielectric Phenomena,

Oct. 16-20, 1983, Buck Hill Falls, PA, pp. 26-34, Oct. 1983.

E. F. Kelley and R. E. Hebner, "Measurement of the Electric-Field in the

Vicinity of an Oil-Pressboard Interface Parallel to the Field," Conf. Record

of 1983 Interfacial Phenomena in Practical Insulating Systems

(Sept. 19-20, 1983), Gaithersburg, Md., pp. 19-22, Dec. 1983.

R. E. Hebner, "Kerr-Effect, Electro-Optical" in Encyclopedia of Physics,

R. G. Lerner and G. L. Trigg, eds., Reading, MA: Add1 son-Wesley, 1981. 483.

R. E. Hebner, "Kerr-Effect, Electro-Optical" in Concise Encyclopedia of Solid

State Physics, R. G. Lerner and G. L. Trigg, eds., Reading, MA:

Addi son-Wesley, 1983. 133.

M. G. Comber, R. Kotter, and R. McKnight, "Experimental Evaluation of

Instruments for Measuring DC Transmission Line Electric Fields and Ion

Currents," IEEE Trans. Power Appar. Syst. PAS-102 , No. 11, pp. 3549-

(Nov. 1983). 725

Ju I
Electronic Instrumentation and Metrology

H. K. Schoenwetter, "An Ultra-Stable AC Power Supply for an Absolute Volt

Determination," Metrologia 10, pp. 11-15 (1974).

R. S. Turgel , "Digital Wattmeter Using a Sampling Method," IEEE Trans.

Instrum. Meas. IM-23 , No. 4, pp. 337-341 (Dec. 1974).

R. S. Turgel, "Sampling Techniques for Electric Power Measurement," Nat. Bur.

Stand. (U.S.), Tech. Note 870, 35 pages (June 1975). [GPO]

H. K. Schoenwetter, "An RMS Digital Voltmeter/Calibrator for Very-Low

Frequencies," IEEE Trans. Instrum. Meas. IM-27 , No. 3, pp. 259-268

(Sept. 1978).

R. S. Turgel and N. M. Oldham, "High-Precision Audio-Frequency Phase

Calibration Standard," IEEE Trans. Instrum. Meas. IM-27 , No. 4,

pp. 460-464 (Dec. 1978).

H. K. Schoenwetter, "NBS Provides Voltage Calibration Service in 0.1 - 10-Hz

Range Using AC Voltmeter/Calibrator," IEEE Trans. Instrum. Meas. IM-28 , No. 4,

pp. 327-331 (Dec. 1979).

A. G. Perrey and H. K. Schoenwetter, "A Schottky Diode Bridge Sampling Gate,"

Nat. Bur. Stand. (U.S.), Tech. Note 1121, 18 pages (May 1980). [GPO]

R. S. Turgel, N. M. Oldham, G. N. Stenbakken, and T. H. Kibalo, "NBS Phase

Angle Calibration Standard," Nat. Bur. Stand. (U.S.), Tech. Note 1144,

143 pages (July 1981). [GPO]

B. A. Bell and 0. Petersons, "ATE Calibration by Means of Dynamic Transport

Standards," IEEE AUTOTESTCON '81 Proceedings, pp. 280-287 (Oct. 1981).

N. M. Oldham and R. S. Turgel, "A Power Factor Standard Using Digital Waveform

Generation," IEEE Trans. Power Appar. Syst. PAS-100 , No. 11, pp. 4435-4438

(Nov. 1981).

J. R. Andrews, N. S. Nahman, and B. A. Bell, "Status of Reference Waveform

Standards Development at NBS," Proceedings of the Waveform Recorder Seminar,

Boulder, CO, October 1981, Nat. Bur. Stand. (U.S.), Spec. Publ. 634,

pp. 69-88 (June 1982). [GPO]

B. A. Bell, B. F. Field, and T. H. Kibalo, "A Fast Response, Low-Frequency

Sampling Voltmeter," Nat. Bur. Stand. (U.S.), Tech. Note 1159, 113 pages

(Aug. 1982), [GPO]

H. K. Schoenwetter, "A Sensitive Analog Comparator," IEEE Trans. Instrum.

Meas. IM-31 , No. 4, pp. 266-269 (December 1982).

N. M. Oldham, "A 50-ppm AC Reference Standard Which Spans 1 Hz to 50 kHz,"

IEEE Trans. Instrum. Meas. IM-32 , No. 1, pp. 176-179 (March 1983).

726
K. J. Lentner and D. R. Flach, "An Automatic System for AC/OC Calibrations,"

IEEE Trans. Instrum. Meas. IM-32 , No. 1, pp. 51-56 (March 1983).

H. K. Schoenwetter, "High Accuracy Settling Time Measurements," IEEE Trans.

Instrum. Meas. IM-32 , No. 1, pp. 22-27 (March 1983).

B. A. Bell and A. G. Perrey, "Peak Conductance Measurements of GaAs Switching

Devices," Picosecond Optoelectronics, Gerard Mourou, Ed., Proc. SPIE 439,

pp. 128-139 (Aug. 1983).

H. K. Schoenwetter, "AC Voltage Calibrations for the 0.1 Hz to 10 Hz Frequency

Range," Nat. Bur. Stand. (U.S.), Tech. Note 1182, 58 pages (Sept. 1983).

[GPO]

H. K. Schoenwetter, "A Programmable Voltage-Step Generator for Testing

Waveform Recorders," IEEE Instrum. and Meas. Technology Conf. Proceed.,

pp. 71-72 (Jan. 1984).

727
Sensors

J. S. Hilten, P. S. Lederer, C. F. Vezzetti, and J. F. Mayo-Wells,

"Development of Dynamic Calibration Methods for Pogo Pressure Transducers,

Nat. Bur. Stand. (U.S.), Tech. Note 927, 58 pages (Nov. 1976). [GPO]

J. S. Hilten, P. S. Lederer, J. F. Mayo-Wells, and C. F. Vezzetti,

"Loose-Particle Detection in Microelectronic Devices," NBSIR 78-1590

(NASA), 73 pages (Jan. 1979). [Limited distribution; not available

from NTIS]

P. S. Lederer, "NBS Interagency Transducer Project 1951 - 1979 — An

Overview," Nat. Bur. Stand. (U.S.), Tech. Note 1110, 45 pages (Aug. 1979).

[GPO]

P. S. Lederer, J. S. Hilten, J. F. Mayo-Wells, and C. F. Vezzetti,

"Loose-Particle Detection in Microelectronic Devices -- A Review of an NBS Task," Proc. Advanced Techniques in Failure Analysis - 1979, pp. 1-19

(Oct. 1979).

J. D. Ramboz, "Measurement and Evaluation Methods for an Angular

Accelerometer," NBSIR 81-2337, 78 pages (Aug. 1981).

P. S. Lederer, "Sensor Handbook for Automatic Test, Monitoring, Diagnostic

and Control Systems Applications to Military Vehicles and Machinery,"

Nat. Bur. Stand. (U.S.), Spec. Publ . 615, 450 pages (Oct. 1981). [GPO]

728
Miscellaneous

A. H. Sher and G. N. Stenbakken, "Selection and Application Guide to

Commercial Intrusion Alarm Systems," Nat. Bur.. Stand. (U.S.), Spec. Publ

.

480-14, 40 pages (Aug. 1979). [GPO]

M. G. Buehler and L. W. Linholm, "Toward a Standard Test Chip Methodology for

Reliable, Custom Integrated Circuits," Proc. of the Custom Integrated Circuit

Conference, pp. 142-146 (May 1981).

M. G. Buehler and D. S. Perloff, "Microelectronic Test Chips and Associated

Parametric Testers: Present and Future," Semiconductor Silicon '81,

H. R. Huff, R. J. Kriegler, and Y. Takeishi, eds., Electrochemical Society,

pp. 859-867 (May 1981).

R. E. Kennerly, R. J. Van Brunt, and A. C. Gallagher, "High-Resolution

Measurement of the Helium Is2s2 2$ Resonance Profile," Physical Review 23,

No. 5, pp. 2430-2442 (May 1981).

T. F. Leedy, G. F. McLane, and G. C. Guenzer, "Transient Temperature

Dependence of Transient Electron Radiation Upset in TTL Nand Gates," IEEE

Trans, on Nuclear Science NS-28 , No. 6 (Dec. 1981).

P. M. Fulcomer, "Field Circuit Breaker Tester," NBSIR 81-2301, 52 pages

(May 1982).

729
Information on Ordering Reports

Publ i cati on Number

GPO

Stock No,

NTIS

Accession No,

NBSIR

98oY*

1 U U /o*

iu <iyo*

in >1 n o in n >1 c 70 /i n o OOM-74-10016

-7/1 C /l A /4-b44 COM-74-11525

/4-bo4 COM-74-11 726

7 C T51 / b- / 0

C0M-75-11071

7 c 7 7/1 /b- / /4 COM-75-11364

7 C 7 7 C /b-/ /b PB 248684

7b-93b PB 299158

/ D- 1 Ub4 PB 25311 0

7 7 1 O 1 1 77- 1 311

7 7 1 "3 1 7

/ /- 1 31

/

PB 274333

78-1590** PB 290679

79-1933 PB 80-119720

79-1950 PB 80-134596

80-1966**

80-2063**

80-2071 PB 80-226699

80-2072 PB 80-197478

80-2177 PB 81-161929

81-2192**

81-2235 PB 81-188005

81-2242 PB 82-134354

81-2267 PB 81-218187

81-2275 PB 81 -216343

81-2283 PB 81-220261

81-2334 PB 81-246860

81-2337 PB 82-115973

81-2360 PB 82-209776

81-2410 PB 82-156993

*Available from author only

**Limited distribution; not available from NTIS.

May be available from author or external

publishing paper.

730
Information on Ordering Reports (cont.)

GPO NTIS

Publ i cation

Number Stock No. Accession No.

NBSIR

81-2301 PB 83-136382

82-2486 PB 82-225723

82-2501 PB 82-227075

82-251 7 PB 82-238353

82-2527 PB 82-263377

82-2528 PB 82-229352

82-2555 PB 83-149187

82-2576 PB 83-180224

82-2586 PB 83-124891

82-2601**

82-2629 PB 83-193110

83-2666 PB 83-192682

83-2705 PB 83-210609

83-2753

83-2755 PB 84-109891

83-2761 PB 84-115104

Technical Note

870 SN003-003-01 434-6 COM- 75- 10946

927 SN003-003-01 683-7 PB 260894

930 SN003-003-01 672-1 PB 258085

1105 PB 80-105331

1121 SN003-003-02197-1 PB 80-198328

1110 SN003-003-02109-1 PB 300-369

1144 SN003-003-02344-2 PB 81-244899

1145 SN003-003-02352-4 PB 81-243404

1155 PB 82-215419

1159 SN003-003-02408-3 PB 83-107011

1179 SN003-003-02505-4 PB 83-252536

1182 SN003-003-02522-4 PB 84-109867

Special Publication

480-14 SN003-003-02098-2 PB 300-604

615 SN003-003-02372-8 PB 82-123746

628 SN003-003-02403-1 PB 82-234121

731
Information on Ordering Reports (cont.)

NTIS

Publication

Number Accession No.

Monograph

109 AD 681-912

Journal of

Research

72C (No. 1 )*

73C (Nos. 3 ft 4)*

75C (Nos. 3 & 4)*

74A (No. 6)*

73C (Nos. 1 & 2)*

Available from author only

732
APPENDIX II

733
NBSIR 84-3014

METROLOGY FOR ELECTROMAGNETIC

TECHNOLOGY:

A BIBLIOGRAPHY OF NBS PUBLICATIONS

Electromagnetic Technology Division

Center for Electronics and Electrical Engineering

National Engineering Laboratory

National Bureau of Standards

U.S. Department of Commerce

Boulder, CO 80303

July 1984

Edited by

R.A. Kamper

K.E. Kline

U.S. DEPARTMENT OF COMMERCE, Malcolm Baldrige, Secretary

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director

734
METROLOGY FOR ELECTROMAGNETIC TECHNOLOGY:

A BIBLIOGRAPHY OF NBS PUBLICATIONS

Edited by

Robert A. Kamper and Kathryn E. Kline

This bibliography lists the publications of the personnel of

the Electromagnetic Technology Division of NBS in the period from

January 1970 through December 1983. A few earlier references that

are directly related to the present work of the Division are incl uded.

Key words: cryoelectronics; electromagnetic metrology; lasers;

microwaves; optical fibers; superconducting materials; time domain

metrology.

INTRODUCTION

The Electromagnetic Technology Division was formed during the reorganization of NBS in April 1978, by combining parts of the former Electromagnetics

and Cryogenics Divisions. It develops measurement methods and standards, and

provides metrological support for: microwave circuits, laser systems, optical

communication equipment, systems using transient or pulsed electromagnetic

phenomena, cryoelectronics, superconductors, and other unusual electrical en- gineering materials. For the individual staff members of the division, the

reorganization brought a realignment of long-term goals but little immediate

discontinuity in their work. It therefore makes good sense that this bibliography should cover a period beginning some time before the reorganization, so as to include at least the more recent origins of the present work of the

division. The editors have attempted to include all work published by the

present staff members of the division, while they were employees of NBS, in

the period from January 1970 to December 1983. There are a few exceptions,

where work that is totally unrelated to the present program has been excluded

or where work by authors now in other parts of NBS has been included because

of its special significance. The work on electromagnetic waveform metrology

was moved to another part of the organization in October 1982. Papers submitted prior to that date are included, but later publications are not. A few

papers on various topics published before 1970 have also been included because

of their direct relationship to the present program.

There are several other sources that may be useful to the reader who is

interested in activities at NBS connected with electromagnetic metrology. A

companion bibliography to this one lists the publications of the Electromagnetic Fields Division. Its topics include metrology for: antennas, satellite communications equipment, electromagnetic interference, and hazard.

Three bibliographies of the publications of the former Electromagnetics Division have been published: NBSIR 73-820 (July 1972-June 1973); NBSIR 74-395

735
(July 1973-June 1974); NBSIR 75-818 (July 1974-June 1975). These were preceded by a series of unpublished reports (edited by H. M. Altschuler) that

cover the period back to 1956. An excellent summary of the whole field of

electromagnetic metrology as it stood in 1967 was published as a special issue

of the IEEE Proceedings (vol. 55; June 1967). Advances in the following

decade were described in another special issue of the same journal (vol. 66;

April 1978).

A Note on Abbreviations

Most readers will be familiar with the commonly used abbreviations for

the names of the various professional journals that appear in this bibliography. There are also some publication series that are peculiar to NBS and

may call for explanation. They are:

NBSIR - NBS Interagency/Internal Report

NBS TN - NBS Technical Note

NBS SP - NBS Special Publication

NBS HB - NBS Handbook

NBS JRES - NBS Journal of Research

NBS MN - NBS Monograph

Purchase Procedures and Document Availability

NBS Technical Notes and Special Publications may be purchased from the

Superintendent of Documents, U.S. Government Printing Office, Washington, DC

20402. Orders must be accompanied by postal money order, express money order,

or check made out to the Superintendent of Documents.

NBS Interagency/Internal Reports (NBSIRs) may be purchased from the

National Technical Information Service, Springfield, VA 22161. Orders must be

accompanied by postal money order, express money order, or check made out to

the NTIS.

Reprints of papers published in non-NBS media may be available in limited

quantities from the authors.

Acknowl edgments

A large part of the labor of preparing a bibliography is spent on collecting and arranging the material. We thank Frances Brown, Edie DeWeese,

Jessie Page, and Sheila Aaker for their assistance with these chores. The

prime source of material was the NTIS file, with access through the Lockheed

Dialog System. This was supplemented with material supplied by the individual

authors

736
MICROWAVE METROLOGY

Millimeter Wave Standards at the National Bureau of Standards (NBS);

Kamper, R. A.; Hoer, C. A.

Proc. SPIE; Aug. 21-25, 1983; San Diego, CA, 423:144-146; 83.

Redundance: A Monitor of Six-Port Performance;

Engen, G. F.

IEE 1983 Colloquium on Advances in S-Parameter Measurement at Microwave

Lengths, May 23, 1983; Savoy Place, London WC2B0BL, p 4/1-4/2; 83.

Choosing Line Lengths for Calibrating Network Analyzers;

Hoer, C.

IEEE/MTT 31(2) :76-78; Jan 83.

Current Trends in NBS Calibration Services;

Kamper, R. A.

NCSL Newsletter, 22(l):38-39; Mar 1982.

Singularities in Calibration of Six-port Network Analyzers;

Ebbesen, H.; Engen, G.

1981 IEEE MTT-S Int. Microwave Symp. Digest, p 149; 81.

WR-10 Single Six-port Measurement System;

Weidman, M.

NBSIR 1650; 81.

A High Power Dual Six-port Automatic Network Analyzer Used in Determining

Biological Effects of rf and Microwave Radiation;

Hoer, C.

IEEE Trans. Microwave Theory Tech., vol. NTT-29; Dec 81.

Control Cable Assemblies Used with Mobile FM Receivers;

Nelson, R. E.; Jickling, R.

National Institute of Justice, NIJ Std. 0216.00; Dec 81.

WR-10 Millimeter Wave Mi crocal orimeter;

Weidman, M.; Hudson, P. A.

NBS TN 1044; Jun 81.

A Least Squares Solution for Use in 6-port Measurement Technique;

Engen, G. F.

IEEE Trans. Microwave Theory Tech., MTT-28(12); Dec 80.

Performance of a Dual Six-Port Automatic Network Analyzer;

Hoer, Cletus A.

Proc. IEEE MTT-S 1979, Int. Microwave Symp. Digest:

The World of Microwaves, Apr 3-May 2, 1979; Orlando, FL, IEEE Cat.

No. 79CHI439-9 MIT-S; 79.

Through-Load-Delay: An Improved Technique for Calibrating the

Dual Six-port Automatic Network Analyzer;

Engen, G. F.; Hoer, C. A.

Int. Microwave Symp. Digest, Orlando, FL, p53; 79.

737
Calibrating a Six-Port Ref 1 ectometer with Four Impedance Standards;

Hoer, Cletus A.

NBS TN 1012; Mar 79.

Advantages of the Six-port Ref lectometer for RF/Microwave Power Measurement

in Operational Systems;

Komarek, E. L. GOMAC Digest; 78.

Calibrating Two Six-Port Ref 1 ectometers With an Unknown Length of

Precision Transmission Line;

Hoer, Cletus A.

Proc. IEEE MIT-S Int. Microwave Symp. Digest, Jun 27-29, 1978; Ottawa,

Canada, pl76-178; 78.

The Application of "Through-Short-Delay" to the Calibration of the

Dual 6-port;

Engen, G. F; Hoer, C. A.; Speciale, R. A.

IEEE Int. Microwave Symp. Digest, pl84-185; 78.

Calibrating the Six-port Refl ectometer •-by Means of Sliding Terminations;

Engen, G. F.

IEEE Trans. Microwave Theory Tech. MTT-26:951-957 ; Dec 78.

NBS RF voltage Comparator;

Driver, L. D.; Ries, F. X.; Rebuldela, G.

NBSIR 78-871; Dec 78.

Status of RF and Microwave Calibration Services at NBS;

Kamper, R. A.

NCSL Newsletter 18:40; Dec 78.

Calibrating Two 6-Port Ref lectometers with Only One Impedance Standard;

Hoer, Cletus A.

NBS TN 1004; Jun 78.

The Six-Port Measurement Technique: A Status Report;

Engen, Glenn F.

Microwave J. 21:18,21-22,24,84,87,89; May 78.

Advances in Microwave Measurement Science;

Engen, Glenn F.

Proc. IEEE 66:374-384; Apr 78.

Instrumentation: Six Ports Simplify Network;

Engen, Glenn F.

Microwave Syst. News, p 54-55; Jan 78.

A Microwave Network Analyzer Using Two 6-Port Refl ectometers

Hoer, Cletus

Proc. 1977 IEEE-MTT-S Int. Microwave Symposium, Jun 21-23, 1977;

San Diego, CA, p 47-49; 77.

738
An Improved Circuit for Implementing the Six-Port Technique of

Microwave Measurements;

Engen, Glenn F.

Proc. 1977 IEEE-MTT-S Int. Microwave Symposium, 21-23 Jun 1977;

San Diego, CA, p 53-55; 77.

An Application of the Six-Port Junction to Precision Measurement of

Microwave One-Port Parameters;

Komarek, E. L. Proc. 1977 IEEE-MTT-S Int. Microwave Symposium, 21-23 Jun 1977;

San Diego, CA, p 56-57; 77.

A Semi -Automated Six-Port for Measuring Millimeter Wave Power and

Complex Reflection Coefficient;

Weidman, Manly P. Proc. 1977 IEEE MTT-S Int. Microwave Symposium, 21-23 Jun 1977;

San Diego, CA, p 58-60; 77.

Design Considerations for Automatic Network Analyzers Based on the

Six-Port Concept;

Engen, Glenn F.

Proc. IEE Euromeas 77 Conf., Sep 5-9, 1977; Sussex, England; London,

England: IEE, p 110-111; 77.

Fixed and Base Station Antennas;

Treado, M. J.; Taggart, H. E.; Nelson, R. E.; Workman, J. L.

NILECJ-STD-0204.00; 77.

The Six-Port Ref lectometer: An Alternative Network Analyzer;

Engen, Glenn F.

Proc. 1977 IEEE-MTT-S Int. Microwave Symposium, 21-23 Jun 1977;

San Diego, CA, p 44-46; 77.

Ultrasonic Calorimeter for Beam Power Measurements from 1 to 15

Megahertz;

Zapf, T. L.; Harvey, M. E.; Larsen, N. T.; Stoltenberg, R. E.

Proc. Ultrasonics Symposium, Sep 29-0ct 1, 1976; Annapolis, MD, IEEE

Cat76 CH1120-5SU, p 573-576; 77.

A Network Analyzer Incorporating Two 6-Port Ref lectometers

;

Hoer, Cletus A.

IEEE Trans. Microwave Theory Tech. MIT-25: 1070-1074; Dec 77.

An Improved Circuit for Implementing the Six-Port Technique of

Microwave Measurements;

Engen, Glenn F.

IEEE Trans. Microwave Theory Tech. MTT-25: 1080-1083; Dec 77.

A Semi -Automated Six-Port for Measuring Millimeter Wave Power and

Complex Reflection Coefficient;

Weidman, Manly P.

IEEE Trans. Microwave Theory Tech. MTT-25: 1083-1085; Dec 77.

739
Performance Characteristics of an Automated Broad-Band Bolometer

Unit Calibration System;

Komarek, Ernest L. IEEE Trans. Microwave Theory MTT-25: 1122-1127 ; Dec 77.

NBS Type IV RF Power Meter Operation and Maintenance;

Larsen, Neil T.

NBSIR 77-866; Oct 77.

Six-Port Measuring Circuit; Patent Application;

Engen, Glenn F.

PAT-APPL-829 381; Filed 31 Aug 77.

The National Electromagnetic Measurement System;

Kamper, R. A.

NBSIR 75-936; Jun 77.

Vector Voltmeter;

Hoer, Cletus A.; Engen, Glenn F.

PATENT-4 001 681; patented 4 Jan 77.

An Automated Broadband System for Measurement of One-Port Microwave

Parameters

Komarek, Ernest L. CPEM Digest, p 167-170; 76.

A New Self-Balancing DC-Substitution RF Power Meter;

Larsen, Neil T.

CPEM Digest, p 203-205; 76.

An NBS Developed Automatic Network Analyzer;

Little, W.; Wakefield, J.; Heim, L.; Allred, C; Zapf, T.

CPEM Digest, p 130-133; 76.

Measurement of Complex Microwave Circuit Parameters Using Only Power

Detectors

Engen, Glenn F.

CPEM Digest, p 171; 76.

Measuring and Minimizing Diode Detector Nonl i neari ty

Hoer, C. A.; Roe, K. C; Allred, C. M.

CPEM Digest, p 108-109; 76.

Reflection Coefficient Standards for Automated Network Analyzers;

Little, W. E.; Yates, B. C.

CPEM Digest, p 128-129; 76.

Repeatability of SMA Coaxial Connectors;

Jesch, R. L. IEEE Trans. Instrum. Meas. IM-25 : 314-320 ; 76.

Scattering Parameters of SMA Coaxial Connectors;

Estin, A. J.

IEEE Trans. Instrum. Meas. IM-25: 329-334 ; 76.

740
A Guide to Voice Scramblers for Law Enforcement Agencies;

Nelson, Robert E.

NBS SP 480-8; Dec 76.

A New Self-Balancing DC-Substitution RF Power Meter;

Larsen, Neil T.

IEEE Trans. Instrum. Meas. IM-25: 343-347 ; Dec 76.

Determination of Microwave Phase and Amplitude from Power

Measurements

;

Engen, Glenn F.

IEEE Trans. Instrum. Meas. IM-25 : 414-418 ; Dec 76.

Ultrasonic Calorimeter for Beam Power Measurements;

Zapf, Thomas L.; Harvey, Morris E.; Larsen, Neil T.; Stoltenberg, Robert E.

NBS TN 686; Sep 76.

A Microwave Vector Voltmeter System;

Roe, Keith C; Hoer, Cletus A.

NBSIR 76-844; Aug 76.

Implementation of the Notch Technique as an RF Peak Pulse Power Standard;

Simpson, Philip A.; Hudson, Paul A.

NBS TN 682; Jul 76.

The Calibration and Use of Directional Couplers Without Standards;

All red, C. McKay; Manney, Charles H. IEEE Trans. Instrum. Meas. IM-25:84-89; Mar 76.

The National Measurement System for Medical Ultrasonics;

Hudson, Paul A.

NBSIR 75-937; Feb 76.

A Precision 30 MHz Waveguide-Below-Cutoff Attenuator with an Absolute

Electronic Readout, NBS Model XII;

Adair, Robert T.

NBSIR 76-833; Jan 76.

Automated Calibration of Di recti onal -Coupl er-Bol ometer-Mount Assemblies;

Engen, Glen F.

Proc. 1975 IEEE Microwave Theory and Technique Symp. Int., May 12-14, 1975;

Palo Alto, CA, p 98-99; 75.

Automated Calibration of Di rectional-Coupler-Bolometer-Mount Assemblies;

Engen, Glenn F.

IEEE Trans. Microwave Theory Tech. MTT-23:984-990; Dec 75.

Using an Arbitrary Six-Port Junction to Measure Complex Voltage Ratios;

Hoer, Cletus A.; Roe, Keith C.

IEEE Trans. Microwave Theory Tech. MTT-23:978-984; Dec 75.

741
Using Six-Port and Eight-Port Junctions to Measure Active and Passive Circuit

Parameters;

Hoer, Cletus A.

NBS TN 673; Sep 75.

Self-Balancing DC-Substitution Measuring System; Patent Application;

Larsen, Neil T.; Reeve, Gerome R.

PAT-APPL-587 565; filed 17 Jun 75.

Characterization of a High Frequency Probe Assembly for Integrated

Circuit Measurements;

Jesch, R> L. ; Hoer, C. A.

NBS TN 663; Apr 75.

An Alternative Calibration Technique for Automated Network Analyzers

with Application to Adapter Evaluation;

Engen, Glenn F. Proc. 1974 IEEE S-MTT Int. Microwave Symposium, Jun 12-14, 1974 Atlanta, GA,

Microwave Symposium Digest, p 261-262; 74.

An Application of the Power Equation Concept and Automation Techniques to

Precision Bolometer Unit Calibration;

Komarek, Ernest L. Proc. 1974 IEEE S-MTT International Microwave Symposium, Jun 12-14, 1974

Atlanta, GA, Microwave Symposium Digest, p 263-265; 74.

An Application of the Power Equation Concept and Automation Techniques to

Precision Bolometer Unit Calibration;

Komarek, Ernest L. ; Tryon, Peter V.

IEEE Trans. Microwave Theory Tech. Part II, 1974 Symposium Issue,

MTT-22: 1260-1267; Dec 74.

Calibration Technique for Automated Network Analyzers with Application to

Adapter Evaluation;

Engen, Glenn F. IEEE Trans. Microwave Theory Tech. Part II, 1974 Symposium Issue,

MTT-22:1255-1260; Dec 74.

Broadband Pulsed/CW Calibration Signal Standard for Field

Intensity Meter (FIM) Receivers;

Simpson, Philip A.

NBSIR 74-371; Jun 74.

Completion of the Program to Evaluate/Improve Instrumentation and Test Methods

for Electroexplosive Device Safety Qualification;

Hudson, Paul A.; Melquist, Dean G.; Ondrejka, Arthur R.; Werner, Paul E. NBSIR 74-379; Jun 74.

Comments on Practical Analysis of Ref lectometers and Power Equation

Concepts;

Engen, Glenn F. IEEE Trans. Instrum. Meas. IM-23: 104-105; Mar 74.

742
Analysis of a Six-Port Junction for Measuring, v, i, a, b, z, Gamma,

and Phase;

Hoer, C. A.; Engen, G. F.

Proc. 6th Conf. Int. Meas. Confederation, Jun 17-23, 1973; Dresden, Germany,

Acta IMEKO 1:213-222; 73.

Sel f-Cal i brati on of Complex Ratio Measuring System;

Allred, C. M.; Manney, C. H.

Proc. Int. Conf. Measurement and Instrumentation, Jun 17-23, 1973; Dresden,

Germany, ACTA IMEKO 1, p 157-166; 73.

Calibration of an Arbitrary Six-Port Junction for Measurement of Active and

Passive Circuit Parameters;

Engen, Glenn F.

IEEE Trans. Instrum. Meas. IM-22: 295-299 ; Dec 73.

Application of a Non-Ideal Sliding Short to Two-Port Loss Measurement;

Weidman, M. P.; Engen, G. F.

NBS TN 644; Oct 73.

Summary of WR15 Flange Evaluation at 60 GHz;

Yates, B. C; Counas, G. J.

NBS TN 642; Oct 73.

Mismatch Considerations in Evaluating Amplifier Noise Performance;

Engen, Glenn F.

IEEE Trans. Instrum. Meas. IM-22 : 274-278 ; Sep 73.

Theory of UHF and Microwave Measurements Using the Power Equation Concept;

Engen, Glenn F.

NBS TN 637; Apr 73.

Application of an 'Arbitrary' 6-Port Junction to Power Measurement Problems;

Engen, Glenn F.; Hoer, Cletus A.

Proc. 1972 Conf. Precision Electromagnetic Measurements, Jun 26-29, 1972,

Boulder, CO; New York, NY: IEEE, p 100-101; 72.

Application of an Arbitrary 6-Port Junction to Power-Measurement

Problems;

Engen, Glenn F.; Hoer, Cletus A.

IEEE Trans. Instrum. Meas. IM-21 :470-474; Nov 72.

The Six-Port Coupler: A New Approach to Measuring Voltage, Current,

Power, Impedance, and Phase;

Hoer, Cletus A.

IEEE Trans. Instrum. Meas. IM-21:466-470; Nov 72.

Broad-Band Lumped-Element Directional Coupler;

Hoer, Cletus A.

PATENT-3 701 057; patented 24 Oct 72.

Mill'imeter Attenuation and Reflection Coefficient Measurement System;

Yates, B. C; Larson, W.

NBS TN 619; Jul 72.

743
The 6-Port Coupler; A New Approach to Measuring V, I, P, Z, and Theta;

Hoer, Cletus A.

Digest of Precision Electromagnetic Measurements Conf., Boulder, CO,

p 26-29; Jun 72.

An Extension to the Sliding Short Method of Connector and

Adaptor Evaluation;

Engen, G. F.

NBS J. Res. 75C: 177-183; 71.

Power Measuring and Leveling System Using a Self-Balancing Bridge;

Larsen, Neil T.; Clague, Frederick R.

PATENT- 3 611 130; patented 5 Oct 71.

An Adjustable-Slot-Length UHF Coaxial Coupler with Decade Bandwidth;

Hudson, P. A.; Saulsbery, L. F.

IEEE Trans. Microwave Theory Tech. MTT-19:781-783; Sep 71.

Standard Field Strength Meter;

Lawton, Robert A.; All red, Charles McKay

PATENT- 3 586 973; patented 22 Jun 71.

International Intercompari son of Power Standards at 3 GHz;

Engen, Glenn F.; Hudson, Paul A.

IEEE Trans. Microwave Theory Tech. MTT-19:411-413; Apr 71.

An Excitation System for Piston Attenuators;

Cook, C. C; Allred, C. M.

IEEE Trans. Instrum. Meas. IM-20: 10-16 ; Feb 71.

Power Equations: A New Concept in the Description and Evaluation of

Microwave Systems;

Engen, Glenn F.

IEEE Trans. Instrum. Meas. IM-20:49-57; Feb 71.

Theorem Giving Limits for S22 When Sll and the Two-Port Efficiency

are Known;

Engen, Glenn F.

IEEE Trans. Instrum. Meas. IM-20:78; Feb 71.

An Immittance Transcomparator;

Agy, David L.; Nelson, R. E.

Annual Precision Measurement Association Metrology Conf. (3rd), Jun 17-18,

1970; Gaithersburg, MD, 1:147-154; 70.

Improvements to the NBS RF Peak-Pulse Power Standard;

Simpson, P. A.; Ondrejka, A. R.

Instrument Society of America Silver Jubilee Int. Conf. and Exhibit,

Oct 1970; Philadelphia, PA, Paper 709-70, p 1-7; 70,

A New Method of Characterizing Amplifier Noise Performance;

Engen, Glenn F.

IEEE Trans. Instrum. Meas. IM-19 : 344-349 ; Nov 70.

744
Broad-Band Resistive-Divider-Type Directional Coupler;

Hoer, C. A.; Agy, D. L.

IEEE Trans. Instrum. Meas. IM-19: 336-343; Nov 70.

A Method for Designing Multi -Screw Waveguide Tuners;

Weidman, M. P.; Campbell, E.

NBS TN 393; Oct 70.

A New Standard for Electric Field Strength;

Lawton, R. A.

IEEE Trans. Instrum. Meas. IM-19:45-51; Feb 70.

A Wide Range CW Power Measurement Technique;

Lawton, R. A.; Allred, C. M.; Hudson, P. A.

IEEE Trans. Instrum. Meas. IM-19:28-34; Feb 70.

An Introduction To The Description and Evaluation of Microwave Systems

Using Terminal Invariant Parameters;

Engen, Glenn F.

NBS MN 112; Oct 69.

Measurement of RF Peak Pulse Power;

Hudson, P. A.

Proc. IEEE 55:851-855; 67.

RF Attenuation;

Russell, D. H.; Larson, W.

Proc. IEEE 55:942-959; Jun 67.

System for Measuring Peak Pulse Power Using Sampling and Comparison

Techniques

;

Hudson, Paul A.; Ecklund, Warner L.; Ondrejka, Arthur R.

PATENT-3 239 758; patented 8 Mar 66.

An Unmodulated Twin-Channel Microwave Attenuation Measurement System;

Russell, D. H.

ISA Trans. 4:162-169; Apr 65.

745
OPTICAL ELECTRONIC METROLOGY

Birefringence Measurements in Single Mode Optical Fiber;

Day, G. W.

Proc. SPIE, August 21-24, 1983, San Diego, CA, 42:72-79; 83.

Fiber Optics: Short-Haul and Long-Haul Measurements and Applications;

Gallawa, R. L. Proc. SPIE, Aug. 24-25, 1982. SPIE, P.O. Box 10, Bellingham, WA 98227;

vol . 355; 83.

Submicrometer Interdigital Silicon Detectors for the Measurement of Picosecond

Optical Pulses;

Phelan, R. J.; Larson, D.; Frederick, N. V.; Franzen, D. L. Proc. SPIE, Single Mode Optical Fibers, 425:207-211; 83.

Estimating Index Profiles of 1.3 ym Single Mode Fibers by Near-Field

Measurements at Blue Wavelengths;

Kim, E. M.; Franzen, D. L. ; Young, M. ; Rodhe, P. M. (Guest Worker)

IEEE J. Lightwave Tech. LT-1(4) :562-566; Dec 83.

Simulating the Scratch Standards for Optical Surfaces—Theory

Johnson, E.

J. Appl. Opt. 22(24) : 4056-4068; Dec 83.

Optical Fiber Characterization Attenuation, Frequency Domain Bandwidth, and

Radiation Patterns;

Chamberlain, G. E.; Day, G. W.; Franzen, D. L.; Gallawa, R. L.; Kim, E. M.;

Young, M.

NBS SP 637, vol. II; Oct. 83.

Linewidth Measurement by High-Pass Filtering- -A New Look;

Young, M.

Appl. Opt. 22(13) :2022-2025; Jul 83.

Laser Measurements;

Sanders, A. A.

Proc. 1983 Measurement Science Conf., Jan. 20-21; Palo Alto, CA; Jun 83.

Measurement of Multimode Optical Fiber Attenuation: An NBS Special Test

Service;

Gallawa, R. L.; Chamberlain, G. E.; Day, G. W. ; Franzen, D. L.; Young, M.

NBS TN 1060; Jun 83.

Optical Time-Domain Refl ectometer Performance and Calibration Studies;

Danielson, B. L. TN-1064; Jun 83.

Two-Dimensional Near-Field Contouring of Optical Fiber Cores;

Kim, E. M.; Franzen, D. L.

Proc. SPIE; Jun 83.

746
Objective Measurements and Characteristics of Scratch Standards;

Young, M.

Proc. SPIE, August 1982, 3526:86-92; Spring 83.

EIA Fiber Performance Measurement Standards;

Gallawa, R.; Franzen, D. L. Photonics Spectra, p 55-68; Apr 83.

Questions Students Ask;

Young, M.

Physics Teacher, p 194-195; Mar 83.

An Inter-Laboratory Measurement Comparison of Core Diameter on Graded- Index

Optical Fibers;

Kim, E. M.; Franzen, D. L. NBS SP 641; Oct 82.

A System for Measuring Energy and Peak Power of Low-Level 1.064 ym Laser

Pul ses;

Sanders, A. A.; Rasumussen, A. L. NBS TN 1058; Oct 82.

Measurement of the Core Diameter of Graded- Index Optical Fibers: An

Interlaboratory Comparison;

Kim, E. M.; Franzen, D. L. Appl. Opt. 21(19) :3443-3450; Oct 82.

Technical Digest - Symposium on Optical Fiber Measurements, 1982;

Franzen, D. L.; Day, G. W.; Gallawa, R. L., eds.

NBS SP 641; Oct 82.

Beam-Profile Measurement of Pulses Using a Spatial Filter to Sample the Hermite Modes for a String of Pulses;

Johnson, E. G.

NBS TN 1057; Sep 82.

Documentation of the NBS C, K, and Q Laser Calibration Systems;

Case W. E.

NBSIR 82-1676; Sep 82.

Optical Fiber Characterization;

Day, G. W.; Danielson, B. L.; Franzen, D. L.; Kim, E. ; Young, M., eds.

NBS SP 637, vol . 1 ; Jul 82.

Calibration Reticle for Optical Fiber Near-Field Core Diameter Measurements;

Kim, E. M.; Franzen, D. L. CPEM Digest; May 82.

Characterization of a Concentric-Core Fiber;

Danielson, B. L.; Franzen, D. L.; Gallawa, R. L.; Kim, E. M.; Young, M.

NBSIR 82-1661; Apr 82.

747
On the Definition of Fiber Numerical Aperture;

Gallawa, R. L. Electro-Optical Systems Design, p 47; Apr 82.

Quantum Noise Limits the Pinspeck Camera to Simple Objects;

Young, M.

J. Opt. Soc. Amer. 72(3) :402-403; Mar 1982.

Optical Waveguide Communications Glossary;

Gallawa, R. L., et al

NBS HB 140; Jan 82.

Book Review: Principles of Optical Fiber Measurements by D. Marcuse;

Young, M.

Laser Focus, p 118-119; Jan 82.

Backscattter Signature Simulations;

Danielson, B. L.

NBS TN 1050; Dec 1981.

Long Optical Fiber Fabry Perot Interferometers;

Franzen, D. L.; Kim, E.

Appl. Opt. 20(23) :3991-3992; 81.

A Measurement Method for Determining the Optical and Electro-optical

Properties of a Thin Film;

Larson, D.

NBSIR 81-1652; Dec 81.

An Optical Waveguide Communications Glossary, Revised;

Hanson, A. G.; Bloom, L. R.; Day, G. W.; Young, M.; Gray, E. M.;

Gallawa, R. L.

NBS HB 140; Dec 81.

Book Review: Optical Fibre Communication (invited review);

Gallawa, R.

IEEE Spectrum, 1 p; Nov 81.

The Use of LEDs as YAG Laser Simulators;

Young, M.

Proc. Conf. Electro-Optics and Lasers; Nov 81.

Optical Fiber Index Profiles by Ref racted-Ray Scanning;

Young, M.

Appl. Opt. 20(19) :3415-21; Oct 81.

Measurement of Optical Fiber Bandwidth in the Frequency Domain;

Day, G. W.

NBS TN 1046; Sep 81.

Standard Measurement Conditions and Test Results on Multimode Fibers;

Franzen, D. L.

Laser Focus, p 103-105; Aug 81.

748
The Use of LEDs to Simulate Weak YAG-laser Beams;

Young, M.

NBS TN 1031; Revised Aug 81.

Interlaboratory Measurement Comparison to Determine the Attenuation and

Bandwidth of Graded-index Fibers;

Franzen, D. L.; Day, G. W.; Danielson, B. L.; Chamberlain, G. E.; Kim, E.

Appl. Opt. 20(14) :2412-2419; 15 Jul 81.

The Characterization of Optical Fiber Waveguides: A Bibliography with

Abstracts, 1970-1980;

Day, G. W.

NBS TN 1043; Jun 81.

Ref racted-ray Scanning (Refracted Near-Field Scanning) for Measuring Index

Profiles of Optical Fibers;

Young, M.

NBS TN 1038; May 81.

Present NBS Capability in Optical Fiber Measurements;

Day, G. W.; Franzen, D. L.

Proc. 1st Int. DoD/Industry Fiber Optics Standards Conf., Washington, DC;

Apr 81.

Results of an Interlaboratory Measurement Comparison Among Fiber Manufacturers

to Determine Attenuation, Bandwidth, and Numerical Aperture of Graded Index

Optical Fibers;

Franzen, D. L.; Day, G. W.; Danielson, B. L.; Kim, E. Conf. Digest 3rd Intl. Conf. Integrated Optics & Optical Fiber Comm.,

San Francisco, CA; Apr 81.

Results of an Inter-Laboratory Measurement Comparison to Determine the

Radiation Angle (NA) of Graded Index Optical Fibers;

Franzen, D. L.; Kim, E.

Appl. Opt. 20(7) :1218-1220; Apr 81.

Sub-nanosecond Electrical Modulation of Light with Hydrogenated Amorphous

Si 1 i con;

Phelan, R. J.; Larson, D. R.; Werner, P. E.

Appl. Phys. Lett. 38(8); Apr 15, 81.

Backscatter Measurements in Optical Fibers;

Danielson, B. L. NBS TN 1034; Feb 81.

Measurement of Far-field and Near-field Radiation Patterns from

Optical Fibers;

Kim, E.; Franzen, D. L. NBS TN 1032; Feb 81.

Progress in Fiber Measurements;

Day, G. W.; Franzen, D. L.

Laser Focus, p 52-56; Feb 81.

749
A System for Characterizing Detectors for the Measurement of Power of CO2

TEA Laser Pulses;

Simpson, P. A.

Proc. 11th Annual Electro-Optical Laser Conf., Oct 23-25, 1979; Ind. and

Sci. Conf. Management, Inc., 222 W. Adams St., Chicago, IL 60606, p 3999-

4007; 80.

The Role of Backscatter Signatures in Optical Fiber Characterization;

Danielson, B. L. Tech. Digest 1st Int. DoD Ind. Fiber Optics Congress, Apr 20, 1980;

Washington, DC; 80.

Linearity and Resolution of Refracted Near-Field Scanning Techniques;

Young, M.

Digest Symposium on Optical Fiber Measurements, NBS SP 597; Oct 80.

Technical Digest, Symposium on Optical Fiber Measurements, 1980;

Day, G. W.; Franzen, D. L., eds.

NBS SP 597; Oct 80.

A System for Measuring the Characteristics of High Peak Power Detectors of

Pulsed C0

2 Laser Radiation;

Simpson, P. A.

NBS TN 1023; Sep 80.

Calibration Technique for Refracted Near-field Scanning of Optical Fibers;

Young, M.

Appl. Opt. 19:2479-80: Aug 80.

Measuring Features of the Fluence at the Far Field of a C0£ Pulsed Laser--

An Issue Study With Suggestions on How To Do It;

Johnson, E. 6, Jr.

NBSIR 80-1628; Apr 80.

An Assessment of the Backscatter Technique as a Means for Estimating Loss

in Optical Waveguides;

Danielson, B. L.

NBS TN 1018; Feb 80.

Measurement of Optical Fiber Bandwidth in the Time Domain;

Franzen, D. L.; Day, G. W.

NBS TN 1019; Feb 80.

LED Source for Determining Optical Detector Time Response at 1.06 Micrometers;

Franzen, D. L.; Day, G. W.

Rev. Sci. Instr. 50:1029; 79.

Linear Systems, Fourier Transforms and Optics by J. D. Gaskill; (book review)

Young, M.

J. Opt. Soc. Amer. 69:637-638; 79.

Measurement of Low Level Laser Pulses at 1.064 ym;

Rasmussen, A. L.; Sanders, A. A.

Measurements of Optical Radiation, SPIE 196:96-103; 79.

750
Measurement of Propagation Constants Related to Material Properties in High

Bandwidth Optical Fibers;

Franzen, D. L.; Day, G. W.,

IEEE J. Quantum Electron. QE-15(12); Dec 79.

Continuous-Wave (Mode-Locked) Dye Laser with Unfolded Cavity;

Young, M.

Appl. Opt. 18:3212; Oct 79.

Limitations Imposed by Material Dispersion on the Measurement of Optical Fiber

Bandwidth with Laser Diode Sources;

Franzen, D. L, ; Day, G. W.,

J. Opt. Soc. Aner. 69(10); Oct 79.

Optical Waveguide Communications Glossary;

Hanson, A. G.; Bloom, L. R.; Day, G. W.; TSallawa, R. L.; Gray, E. M.;

M. Young

NTIA SP 79-4; Sep 79.

Time Domain Pulse Measurements and Computed Frequency Response of Optical

Communications Components;

Andrews, J. R.; Young, M.

NBSIR 79-1620; Sep 79.

National Standards of a Powerful Sort;

Sanders, A. A.

Opt. Spectra 13:45; Aug 79.

Quality Assurance Program for the NBS, C, K, and Q Laser Calibration Systems;

Case, W. E.

NBSIR 79-1619; Aug 79.

Design of a Reflection Apparatus for Laser Beam Profile Measurements;

Johnson, E. G., Jr.

NBS TN 1015; Jul 79.

Fiber Measurements: Quality and Cost;

Day, G. W.

Proc. Int. Conf. on Communications, Boston; Jun 79.

Attenuation Measurements on Optical Fiber Waveguides: An Interl aboratory

Comparison Among Manufacturers;

Day, G. W.; Chamberlain, G. E.

NBSIR 79-1608; May 79.

Conference on Optical Scattering Standards;

Young, M.

Proc. SPIE Conf.; Apr 79.

Laser Beam Profile Measurements Using Spatial Sampling, Fourier

Optics, and Holography;

Johnson, Eric G., Jr.

NBS TN 1009; Jan 79.

751
A Simple First Positive System Nitrogen Laser for use in Optical

Fiber Measurements;

Franzen, D. L.; Danielson, B. L.; Day, G. W

IEEE J. Quantum Electron. QE-1 4:548; 78.

Measurement Problems in Multimode Optical Waveguides;

Day, G. W.

Proc. Int. Com. Conf., Jun 1978; Toronto, Canada, vol. 1; 78.

Optical Fiber Phase Discriminator;

Danielson, B. L. Appl. Opt. 17:3665-3668; Nov 78.

Improvements in a Calorimeter for High-Power CW Lasers;

Chamberlain, George E.; Simpson, Philip A.; Smith, Richard L.

IEEE Trans. Instrum. Meas. IM-27 :81-86 ; Mar 78.

Laser Far-Field Beam-Profile Measurements by the Focal Plane Technique;

Day, G. W.; Stubenrauch, C. F.

NBS TN 1001; Mar 1978.

Fiber Optics Metrology at NBS;

Danielson, B.; Day, G.; Franzen, D.

Proc. Union Radio Scientifique Internationale Comm. Symp., Oct 3-7, 1977;

Lannion, France, (International Union of Radio Science, France), p 430-431;

Dec 77.

Evaluating the Inequivalence and a Computational Simplification for

the NBS Laser Energy Standards;

Johnson, Eric G., Jr.

Appl. Opt. 16:2315-2321; Aug 77.

Measurement Procedures for the Optical Beam Splitter Attenuation

Device BA-1;

Danielson, B. L. NBSIR 77-858; May 77.

Proposed Standards for Ladar Signatures;

Danielson, B. L.

NBSIR 77-856; Apr 77.

An NBS Laser Measurement Assurance Program (MAP);

Sanders, A. A.; Cook, A. R.

Proc. Electro-Optical Systems Design Conf. 1976 and Int. Laser Exposition,

Sep 14-16, 1976; New York, p 277-280; 76.

Laser Action in Sputtered Metal Vapors;

McNeil, J. R.; Johnson, W. L.; Collins, G. J.; Persson, K. B.;

Franzen, D. L.

9th Int. Conf. on Quantum Electronics, Jun 14-16, 1976; Amsterdam, The

Netherlands, p 162-163; 76.

752
Performance and Characteristics of Polyvi nyl i dene Fluoride

Pyroelectric Detectors;

Day, G. W.; Hamilton, C. A.; Gruzensky, P. M.; Phelan, R. J., Jr.

Proc IEEE Symposium on Application of Ferroelectrics, Jun 9-11, 1975;

Ai; querque, NM; Ferroelectrics 10:99-102; 76.

Absolute Reference Calorimeter for Measuring High Power Laser Pulses;

Franzen, D. L.; Schmidt, L. B.

Appl. Opt. 15:3115-3122; Dec 76.

Spectral Reference Detector for the Visible to 12 Micrometer Region;

Convenient, Spectrally Flat;

Day, G. W.; Hamilton, C. A.; Pyatt, K. W.

Appl. Opt. 15:1865-1868; Jul 76.

An Electrically Calibrated Pyroelectric Radiometer System;

Hamilton, C. A.; Day, G. W.; Phelan, R. J., Jr.

NBS TN 678; Mar 76.

Ultraviolet Laser Action from Cu II in the 2500-A Region;

McNeil, J. R.; Collins, G. J.; Persson, K. B.; Franzen, D. L.

Appl. Phys. Lett. 28:207-209; Feb 76.

Laser Attenuators for the Production of Low Power Beams in the Visible and

1.06 micron Regions;

Danielson, B. L.; Beers, Yardley

NBS TN 677; Jan 76.

Radiometry Without Standard Sources/Electrically Calibrated Pyroel ectri cs

;

Phelan, Robert J. Jr; Hamilton, Clark A.; Day, Gordon W.

Proc. Society of Photo-Optical Instrumentation Engineers, Aug 17-22, 1975;

San Diego, CA, Paper in Modern Utilization of Infrared Technology Civilian

and Military, SPIE 62:159-165; 75.

Pyroelectric Radiometers Get Off The Drawing Board;

Hamilton, C. A.; Phelan, R. J., Jr; Day, G. W.

Opt. Spectra 9:37-38; Oct 75.

Improving Beam Measurement;

Smith, R. L.; Sanders, A. A.

Laser Focus, p 70-71; Apr 75.

Precision Beam Splitters for C02 Lasers;

Franzen, Douglas L. Appl. Opt. 14:647-652; Mar 75.

A Pyroelectric Power Meter for the Measurement of Low Level Laser

Radi ati on; Hamilton, C. A.; Day, G. W.

NBS TN 665; Feb 75.

The Polarization of PVF and PVF2 Pyroel ectri cs

;

Phelan, R. J. Jr; Peterson, R. L.; Hamilton, C. A.; Day, G. W.

Ferroelectrics 7:375-377; 74.

753
Current Status of NBS Low-Power Laser Energy Measurement;

West, E. D. ; Case, W. E. Proc. Conf. on Precision Electromagnetic Measurements, Jul 1-5, 1974;

London, England, IEEE Trans. Inst. Meas. IM-23 : 422-425 ; Dec 74.

Analysis of Response of Pyroelectric Optical Detectors;

Peterson, R. L.; Day, G. W.; Gruzensky, P. M.; Phelan, R. J.

J. Appl. Phys. 45:3296-3303; Aug 74.

Effects of Poling Conditions on Responsivity and Uniformity of Polarization

of PVF2 Pyroelectric Detectors;

Day, G. W.; Hamilton, C. A.; Peterson, R. L.; Phelan, R. J., Jr;

Mullen, L. 0.

Appl. Phys. Lett. 24:456-458; May 74.

Absolute, Pyroelectric Radiometers and Two Dimensional Arrays;

Phelan, R. J., Jr; Peterson, R. L.; Klein, G. P.; Hamilton, C. A.;

Day, G. W.

Proc. American Electro-Optical Systems Design Conf., New York, NY,

p 117-123; 73.

Comparison of the Laser Power and Total Irradiance Scales Maintained by the

National Bureau of Standards;

Geist, Jon; Schmidt, L. B.; Case, W. E.

Appl. Opt. 12:2773-2776; Nov 73.

Electrically Calibrated Pyroelectric Optical-Radiation Detector;

Phelan, Robert J., Jr; Cook, A. R.

Appl. Opt. 10:2494-2500; Oct 73.

Continuous Laser-Sustained Plasmas;

Franzen, D. L.

J. Appl. Phys. 44:1727-1732; Apr 73.

Limitations of the Use of Vacuum Photodiodes in Instruments for the

Measurement of Laser Power and Energy;

Smith, R. L.; Phelan, R. J., Jr.

Appl. Opt. 12:795-798; Apr 73.

Accurate Frequencies of Molecular Transitions Used in Laser Stabilization:

The 3.39-mi crometer Transition in CH4 and the 9.33- and 10 .18-mi crometer

Transitions in C02;

Evenson, K. M.; Wells, J. S.; Peterson, F. R.; Danielson, B. L.; Day, G. W

Appl. Phys. Lett. 22:192-195; Feb 73.

A Calorimeter for High Power CW Lasers;

Smith, R. L.; Case, W. E.; Rasmussen, A. L.; Russell, T. W.; West, E. D.

Proc. Conf. on Precision Electromagnetic Measurements, Jun 26-29, 1972;

Boulder, CO, p 138-139; 72.

High D*, Fast, Lead Zirconate Titanate Pyroeleci- : c Detectors;

Mahler, R. J.; Phelan, R. J., Jr; Cook, A. R.

Infrared Phys. 12:57-59; 72.

754
A Calorimeter for High-Power CW Lasers;

Smith, R. L.; Russell, T. W.; Case, W. E.; Rasmussen, A. L.

IEEE Trans. Instrum. Meas. IM-21:434-8; Nov 72.

Speed of Light from Direct Frequency and Wavelength Measurements of

the Methane-Stabilized Laser;

Evenson, K. M.; Wells, J. S.; Petersen, F. R.; Danielson, B. L.;

Day, G. W.

Phys. Rev. Lett. 29:1346-1349; Nov 72.

CW Gas Breakdown in Argon Using 10.6 Micrometer Laser Radiation;

Franzen, Douglas L. Appl. Phys. Lett. 21:62-64; Jul 72.

Role of Infrared Frequency Synthesis >n Metrology;

Wells, J. S.; Evenson, K. M.; Day, G. W.^ Halford, Donald

Proc. IEEE Lett. 60:621-623; May 72.

Gain Saturation Measurements in C02, TEA Amplifiers;

Franzen, Douglas L.; Jennings, Donald A.

J. Appl. Phys. 43:729-730; Feb 72.

Double Plate Calorimeter for Measuring the Reflectivity of the

Plates and the Energy in a Beam of Radiation;

Rasmussen, Alvin L. PATENT-3 622 245; patented 23 Nov 71.

High D* Pyroelectric Polyvinylfluoride Detectors;

Phelan, Robert J., Jr; Mahler, Robert J.; Cook, Alan R.

Appl. Phys. Lett. 19:337-338; Nov 71.

Measurement of Laser Energy of Linear Components of Polarization at

1.060 micron;

Rasmussen, A. L.

Rev. Sci. Instrum. 42:1590-1593; Nov 71.

Linear and Nonlinear Optical Properties of Trigonal Selenium;

Day, G. W.

Appl. Phys. Lett. 18:347-349; Apr 71.

Extension of Absolute Frequency Measurements to the cw He-Ne Laser

at 88 THz (3.39 micrometer);

Evenson, K. M. ; Day, G. W. ; Wells, J. S. ; Mullen, L. 0.

Appl. Phys. Lett. 20:133-134; Feb 71.

Some Optical Properties of Cesium Cupric Chloride;

Day, G. W.; Gruzensky, P. M.

Appl. Opt. 9:2494-2795; 70.

Laser Power and Energy Measurements;

Jennings, D. A.; West, E. D.; Evenson, K. M.; Rasmussen, A. L.;

Simmons, W. R.

NBS TN 382; Oct 69.

755
InSb-GaAsP Infrared to Visible Light Converter;

Phelan, Robert J., Jr.

Proc. IEEE 55:1505-1502; Aug 67.

InSb MOS Infrared Detector;

Phelan, R. J., Jr.; Dimmock, J. 0.

Appl. Phys. Lett. 10:55-7; Jan 67.

Incoherent Source Optical Pumping of Visible and Infrared

Semiconductor Lasers;

Phelan, Robert J., Jr.

Proc. IEEE 54:1119-20; Aug 66.

Laser Emission by Optical Pumping of Semiconductors;

Phelan, R. J., Jr.

Proc. Quantum Electronics Conf., p 435-441; Jul 65.

756
ELECTROMAGNETIC WAVEFORM METROLOGY

Reference Waveform Flat Pulse Generator;

Andrews, J.; Bell, B.; Nahman, N. S.; Baldwin, E. E. Proc. CPEM 1982 in IEEE Trans. Instrum. Meas. IM-32(1) :27-32; Mar 83.

Reference Waveform Flat Pulse Generator;

Andrews, J. R.; Bell, B. A.; Nahman, N. S.; Baldwin, E. E.

1982 CPEM Conf. Digest; Jun 82.

Deconvol ution of Time Domain Waveforms in the Presence of Noise;

Nahman, N. S.; Guillaume, M. E. NBS TN 1047; Oct 81

.

Shielded Balanced and Coaxial Transmission Lines - Parametric Measurements and

Instrumentation Relevant to Signal Waveform Transmission in Digital Service;

Gans, W. L.; Nahman, N. S.

NBSIR 81-1042; Jun 81.

Amplitude Calibrator for Oscilloscopes;

Andrews, J. R.; Baldwin, E. E. NBSIR 81-1646; Apr 81.

Error Criteria and the Use of Reference Waveforms;

Nahman, N. S.

Electron. Test 4(2):72-76; Feb 81.

Instrument Modeling and Deconvol ution Improve Time Domain Calibrations;

Nahman, N. S.; Andrews, J. R. NBS Dimensions, p 21-23; Jan/Feb 81.

Deconvol ution of Duration Limited Waveforms;

Nahman, N. S.; Guillaume, M. E.

Digest of Papers, URSI/U.S. National Radio Science Meeting, Boulder, CO,

p 86; Jan 81.

Error Criteria and the Use of Reference Waveforms;

Nahman, N. S.

Proc. ATE Seminar/Exhibit, Pasadena, CA, Jan 1981, Benwill Publications,

Boston, MA, p IV.21-IV.28; Jan 81.

Pulse Reference Waveform Standards Development at NBS;

Andrews, J. R. Proc. ATE Seminar/Exhibit, Pasadena, CA, Jan 1981, Benwill Publications,

Boston, MA, p IV.13-IV.19; Jan 81.

The Estimation of the Pulse Waveform in the Calibration of Impulse Generators;

Andrews, J. R.

Digest of Papers, URSI/U.S. National Radio Science Meeting, Boulder, CO,

p 87; Jan 81.

757
Applications of Time Domain Methods to Microwave Measurements;

Nahman, N. S.; Andrews, J. R.; Gans, W. L.; Guillaume, M. E.; Lawton, R. Ondrejka, A. R. ; Young, M.

IEE Proc. (Microwaves, Optics, and Acoustics) Special Issue on Microwave

Measurements 127, Part H(2) :171-178; Apr 80.

A Superconducting Sampler for Josephson Logic Circuits;

Hamilton, C; Lloyd, F.; Peterson, R.; Andrews, J. R.

Appl. Phys. Lett. 35(9) :718-719; Nov 79.

A Video Recorder for Coherent Doppler Radar;

Wharton, M. J.; Frush, C. L. ; Nahman, N. S.

IEEE Trans. Geosci . Electron., Special Issue on Radio Meteorology,

GE-17C4) :171-178; Oct 79.

Time Domain Pulse Measurements and Computed Frequency Responses of Optical

Communications Components;

Andrews, J. R., Young, M.

NBSIR 1620; Sep 79.

Radar Absorber Measurement Techniques at Frequencies above 20 GHz;

Nahman, N. S.; All red, C. M.; Andrews, J. R.; Hoer, C. A.; Lawton, R. A.

NBSIR 1613; Aug 79.

Picosecond Domain Waveform Measurements (Japanese translation);

Nahman, Norris S.

Nikkei Electronics, Tokyo Japan: Nikkei -McGraw Hill, Inc.; Apr 79,

Measurement of Pulsed-Laser Power;

Young, M. ; Lawton, R. A.

NBS TN 1010; Feb 79.

Antennas and the Associated Time Domain Range for the Measurement of

Impulsive Fields;

Lawton, Robert A.; Ondrejka, Arthur R.

NBS TN 1008; Nov 78.

Applications of the Homomorphic Transformation to Time Domain

Measurement Problems;

Riad, Sedki M.; Nahman, Norris S.

NBSIR 78-881; Jun 78.

SHF Impulse Generator;

Andrews, James R.; Baldwin, Eugene E.

NBSIR 78-888; Jun 78.

Broadband Isotropic Antenna with Fiber-Optic Link to a Conventional

Receiver;

Larsen, Ezra B.; Andrews, James R.; Baldwin, Eugene E.

PATENT-4 091 327; patented 23 May 78.

Automatic Network Measurements in the Time Domain;

Andrews, James R. Proc. IEEE 66:414-423; Apr 78.

758
>

Picosecond-Domain Waveform Measurements;

Nahman, Norn's S.

Proc. IEEE 66, 441-454; Apr 78.

Saturation of Silicon Photodiodes at High Modulation Frequency;

Young, M.; Lawton, R. A.

Appl. Opt. 17:1103-1106; 1 Apr 78.

Measurement on Pulses and Pulse Transmission Media, Circuits and Components;

Andrews, J. R. URSI Review of Radio Science, p 9; 1975-1977.

Application of a Automated Pulse Measurement System To Telecommunications

Measurements

;

Gans, W.; Andrews, J. R.; Riad, S.; Cozannet, A.; Debeau, J.

Collogue International sur la Measure en Telecommunications, URSI, CNET, Oct

3-7, 1977; Lannion, France, p 165-170; 77'.

Saturation of Optical Detectors at High Modulation Frequency;

Young, M.; Lawton, R. A.

Proc. Annual Meeting Optical Society of America, Oct 12, 1977; Toronto,

Canada, J. Opt. Soc. Am. 67:1398; 77.

Signal Waveform Metrology at NBS;

Nahman, Norris S. Proc. WESCON/77, Electronics at the Golden Gate, Sep 19-21, 1977; San

Francisco, CA; 77.

Standards for the Measurement of Impulsive Fields Radiated by a TEM Horn

Antenna;

Lawton, Robert A.; Ondrejka, Arthur R. Proc. Union Radio Scientifique International Comm. Symp., Oct 3-7, 1977;

Lannion, France, p 1-4; Dec 77.

The Measurement of Pulse Transition Duration;

Andrews, James R.; Nahman, Norris S.

Proc. Union Radio Scientifique Internationale Comm. Symp., Oct 3-7, 1977;

Lannion, France, p 159-164; Dec 77.

Laser-Mode Beating Used for Detector Frequency-Response Measurements;

Lawton, R. A.; Young, M.

Appl. Opt. 16:2703-2705; Oct 77.

Waveform Sampler;

Lawton, Robert A.; Andrews, James R.

PATENT-4 030 840; patented 21 Jun 77.

Application of the Homomorphic Deconvoluti on for the Separation of

TDR Signals Occurring in Overlapping Time Windows;

Riad, Sedki M.; Nahman, Norris S.

CPEM Digest, p 27-29; 76.

759
Impulse Generator Spectrum Amplitude Measurement Techniques;

Andrews, James R. CPEM Digest, p 23-24; 76.

Present Capabilities of the NBS Automatic Pulse Measurement System;

Gans, William L. CPEM Digest, p 25-26; 76.

Using Fiber Optics in a Broadband, Sensitive, Isotropic Antenna - 15 kHz to 150 MHz;

Larsen, E. B.; Andrews, J. R. Proc. IEEE International Symposium on Electromagnetic Compatabil i ty

,

Jul 13-15 1976; Washington, DC, p 385-389; 76.

Application of the Homomorphic Deconvolution for the Separation of

TDR Signals Occurring in Overlapping Time Windows;

Riad, Sedki M.; Nahman, Norris S.

IEEE Trans. Instrum. Meas. IM-25 :388-391; Dec 76.

Impulse Generator Spectrum Amplitude Measurement Techniques;

Andrews, James R. IEEE Trans. Instrum. Meas. IM-25:380-384; Dec 76.

Present Capabilities of the NBS Automatic Pulse Measurement System;

Gans, William L. IEEE Trans. Instrum. Meas. IM-25:384-388; Dec 76.

Electrical Strobing of a Photoconductor Cuts Sampling Oscilloscope's

Ri setime;

Lawton, Robert A.; Andrews, James R.

Laser Focus, p 62-65; Nov 76.

Sensitive Isotropic Antenna with Fiber-Optic Link to a Conventional

Receiver;

Larsen, E. B.; Andrews, J. R.; Baldwin, E. E. NBSIR 75-819; Sep 76.

Andrews, James R.

PATENT- 3 967 140; patented 29 Jun 76.

Electrically Strobed Optical Waveform Sampling Oscilloscope;

Andrews, James R.; Lawton, Robert A.

Rev. Sci. Instrum. 47:311-313; Mar 76.

Optically Strobed Sampling Oscilloscope;

Lawton, Robert A.; Andrews, James R. IEEE Trans. Instrum. Meas. 25:56-60; Mar 76.

Baseband Impulse Generator Useful to 5 GHz;

Andrews, James R.; Baldwin, Eugene E.

Proc. 1975 IEEE Int. Symp. on Electromagnetic Compatibility, Oct 7-9, 1975;

San Antonio, TX, p 1-4; 75.

760
Picosecond Pulse Research at NBS;

Andrews, James R.; Lawton, Robert A.

Proc. Joint Measurement Conf., Nov 12-14, 1974; Gai thersburg, MD, p 123-140;

75.

Time Domain Automatic Network Analyzer;

Andrews, J. R.; Gans, William L. Proc. Colloque International sur L' Electronique et la Mesure, May 26-30,

1975; Paris, France, La Mesure Electriquede Precision, p 258-267; 75.

Time Domain Automatic Network Analyzer;

Andrews, J. R.; Gans, W. L. L'Onde Electrique, Paris, France, 55( 10) :569-574; Dec 75.

Di recti onal -Coupler Technique for Triggering a Tunnel Diode;

Andrews, James R. IEEE Trans. Instrum. Meas. IM-24: 275-277 ; Sep 75.

Time Domain Automatic Network Analyzer for Measurement of RF and

Microwave Components;

Gans, William L.; Andrews, James R.

NBS TN 672; Sep 75.

Pulsed-Laser Application to Sampling Oscilloscope;

Lawton, Robert A.; Andrews, James R. Electron. Lett. 11:138; Apr 75.

Pulsed Wavemeter Timing Reference for Sampling Oscilloscope

Cal ibration;

Andrews, James R.; Gans, William L. IEEE Trans. Instrum. Meas. IM-24:82; Mar 75.

Photoconductive Detector of Fast-Transition Optical Waveforms;

Lawton, R. A.; Scavannec, A.

Electron. Lett. 11:74-75; Feb 75.

Application of Ultrashort Optical Pulses to Electrical Pulse Measurements;

Honda, T.; Nahman, N. S.

Proc. Symposium on New Instrumentation, Jan 28-29, 1974; IEE of Japan,

Hokkaido University, p 29-38; 74.

Precision Picosecond-Pulse Measurements Using a High Quality

Superconducting Delay Line;

Andrews, James R.

Proc. Conf. on Precision Electromagnetic Measurements, Jul 1-5, 1974;

London, England; Abstract in CPEM Digest, London: IEE; p 316-318; 74.

Precision Picosecond-Pulse Measurements Using a High-Quality

Superconducting Delay Line;

Andrews, James R.

IEEE Trans. Instrum. Meas. IM-23:468-472 ; Dec 74.

761
Picosecond Pulse Generators Using Microminiature Mercury Switches;

Andrews, James R. NBSIR 74-377; Mar 74.

Active and Passive Mode Locking of Continuously Operating Rhodamine

6G Dye Lasers;

Scavennec, Andre; Nahman, N. S.

NBSIR 73-347; Feb 74.

Inexpensive Laser Diode Pulse Generator for Optical Waveguide

Studies;

Andrews, James R. Rev. Sci. Instrum. 45:22-24; Jan 74.

A Simple Passively Mode-Locked CW Dye Laser;

Scavennec, Andre; Nahman, Norris S.

IEEE J. Quantum Electron. QE-10:95-96; Jan 74.

Autocorrelation and Power Measurement with Pyroelectric and

Dielectric Bolometers;

Lawton, Robert A.

IEEE Trans. Instrum. Meas. IM-22 : 299-306 ; Dec 73.

Random Sampling Oscilloscope for the Observation of Mercury Switch

Closure Transition Times;

Andrews, James R.

IEEE Trans. Instrum. Meas. IM-22 : 375-381 ; Dec 73.

Frequency Domain Measurement of Baseband Instrumentation;

Nahman, N. S.; Jickling, R. M.

NBSIR 73-330; Jul 73.

Pulse Testing of RF and Microwave Components;

Gans, William L.; Nahman, N. S.

NBSIR 73-325; Jul 73.

Random Sampling Oscilloscope Time Base;

Andrews, James R. NBSIR 73-309; Jun 73.

A Note on the Transition (Rise) Time Versus Line Length in Coaxial

Cables;

Nahman, Norris S.

IEEE Trans. Circuit Theory CT-20: 165-167 ; Mar 73.

Miniature Superconductive Coaxial Transmission Lines;

Nahman, Norris S.

Proc. IEEE 61:76-79; Jan 73.

Coaxial Line Pulse-Response Error Due to a Planar Skin Effect

Approximati on; Holt, D. R. ; Nahman, N. S.

Proc. 1972 Conf. on Precision Electromagnetic Measurement, Jun 26-29, 1972;

Boulder, CO; p 1-2; 72.

762
Fast Fourier Transform Implementation for the Calculation of Network

Frequency Domain Transfer Functions from Time Domain Waveforms;

Gans, William L.; Nahman, N. S.

NBSIR 73-303; Dec 72.

Reference-Waveform Generation Using Debye Dielectric Dispersion;

Nahman, N. S.; Jickling, R. M. ; Holt, D. R. NBSIR 73-304; Dec 72.

Coaxial-Line Pulse-Response Error Due to a Planar Skin-Effect

Approximation;

Holt, D. R.; Nahman, N. S.

IEEE Trans. Instrum. Meas. IM-21 :515-519; Nov 72.

A Dye Model ocked Nd+3 Glass Laser for- Generating Electrical Reference

Waveforms;

Honda, T.; Nahman, N. S.

NBSIR 74-360; Sep 72.

Transient Analysis of Coaxial Cables Using the Skin Effect Approximation

A + B(the square root of s);

Nahman, N. S.; Holt, D. R. IEEE Trans. Circuit Theory CT-19:443-451; Sep 72.

Measurement of Laser Pulse Dispersion in Glass Fiber Optical Waveguides;

Andrews, J. R. NBS Memo for the Record; Jul 72.

Measurement of Broadening of Pulses in Glass Fibers;

Bouille, R.; Andrews, J. R. Electron. Lett. 8:309-310; Jun 72.

Pyroelectric Detector Application to Baseband Pulse Measurements;

Lawton, Robert A.; Nahman, N. S. Electron. Lett. 8:318-320; Jun 72.

A Frequency Calibrator for UHF Using an Avalanche Transistor;

Andrews, James R. Q S T 56:16-18; May 72.

Survey of Present Waveform Sampling System Limitations;

Andrews, J. R.; Whittemore, T.; McCaa, W. D. NBS Report 10-731; Feb 72.

Electron-Beam Deflection in Traveling-Wave Oscilloscopes;

Andrews, James R.; Nahman, N. S. Record of Symposium on Electron, Ion, and Laser Beam Technology (11th),

May 12-14, 1971; Boulder, CO; 71.

An Interfacing Unit for a 28 psec Feedthrough Sampling Oscilloscope;

Andrews, J. R. Rev. Sci. Instrum. 42:1882-1885; Dec 71.

763
Horizontal Time Base Sweep Generator for a Traveling Wave Oscilloscope;

Andrews, James R. IEEE Trans. Nucl . Sci. NS-18:3-8; Oct 71.

The Measured Time and Frequency Response of a Miniature Superconducting

Coaxial Line;

Ekstrom, M. P.; McCaa, W. D. Jr; Nahman, N. S.

IEEE Trans. Nucl. Sci. NS-18:18-25; Oct 71

Generation of Reference Waveforms by Uniform Lossy Transmission Lines;

McCaa, W. L).; Nahman, N. S.

IEEE Trans. Instrum. Meas. IM-19 : 382-390 ; Nov 70.

Improved Bias Supply for Tunnel -Diode Picosecond Pulse Generators;

Andrews, J. R.

IEEE Trans. Instrum. Meas. IM-19 : 171-175 ; Aug 70.

Measured Time and Frequency Response of a Miniature Superconducting

Coaxial Line;

Nahman, N. S.; Ekstrom, M. P.; McCaa, W. D., Jr.

Applied Superconductivity Conf., Boulder, CO; Jun 70.

Deflection Theory of Traveling Wave Oscilloscopes;

Andrews, J. R. PhD Dissertation, Univ. of Kansas; Apr 1970. Also NBS Memo for the Record.

Tunnel Diode Pulse Generator and Bias Supply;

Andrews, J. R. NBS Memo for the Record; Aug 69.

Frequency and Time Domain Analysis of A Superconductive Coaxial Line

Using the BCS Theory;

McCaa, W. D.; Nahman, N. S.

J. Appl. Phys. 40:2098-2099; Apr 69.

A Digital Computer Technique for Calculating the Step Response of Lumped

or Distributed Networks;

Andrews, J. R.; Nahman, N. S.

NBS JRES, Sec. B, p 163-176; April-June 69.

Investigation and Evaluation of Methods for Pulse Rise-Time Measurement;

Hudson, P.; Andrews, J. R.; Blank, W.; Ondrejka, A.

Tech Report RADC-TR-68-207 ; Oct 68.

The Measurement of Baseband Pulse Rise-Times of Less than 10" 9 Seconds;

Nahman, N. S.

Proc. IEEE 55:855-864; Jun 67.

Random Sampling Osci 11 ography

Frye, G. J., and Nahman, N. S.

IEEE Trans. Instrum. Meas. IM-13 :8-13; Mar 64.

764
CRYOELECTRONIC METROLOGY

Cryogeni cs

;

Zimmerman, J. E.

Chapt. 3 in Bi omagneti sm: an Interdisciplinary Approach, Samuel J.

Williamson, et al . , eds., NATO Advanced Study Institute Publication,

vol. 66, p 43-67, New York: Plenum Press; 83.

Magnetic Quantities, Units, Materials, and Measurements;

Zimmerman, J. E.

Chapt. 2 in Bi omagnetism: an Interdisciplinary Approach, Samuel J.

Williamson, et al., eds., NATO Advanced Study Institute Publication,

vol. 66, p. 17-42, New York: Plenum Press; 83.

A Cryocooler for Applications Requiring Low Magnetic and Mechanical

Interference;

Zimmerman, J. E.; Daney, D. E.; Sullivan, D. B.

NASA Pub. 2287, p 95-106; Dec 83.

An Approach to Optimization of Low-Power Stirling Cryocoolers;

Sullivan, D. B.

Proc. 2nd Biennial Conf. Refrigeration for Cryogenic Sensors; Dec. 7-8,

1982, Greenbelt, MD; NASA Conf. Publ. 2287, p 95-106; Dec 83.

Summary of the Proceedings of the 2nd Biennial Conf. Refrigeration for

Cryogenic Sensors;

Zimmerman, J. E.

Cryogenics 23(5) :281-282; May 1983.

Chaos in Josephson Circuits;

Kautz, R. L.

Proc. Applied Superconductivity Conf., IEEE Trans. Magn. MAG-19(3) :465-473;

May 83.

Double Transformer Coupling to a Very Low Noise SQUID;

Cromar, M.; Muhlfelder, B.

Proc. Applied Superconductivity Conf., IEEE Trans. Magn. MAG-19(3) :303-307;

May 83.

8-Bit Superconducting A/D Converter;

Hamilton, C. A.; Lloyd, F. L. Proc. Applied Superconductivity Conf., IEEE Trans. Magn. MAG-19( 3) 1259-1261

;

May 83.

Microwave Mixing and Direct Detection Using SIS and SIS' Quasiparticle Tunnel

Juncti ons

;

Harris, R. E., et al

.

Proc. Applied Superconductivity Conf., IEEE Trans. Magn. MAG-19(3) :490-493;

May 83.

100 GHz Binary Counter Using SQUID Flip Flops;

Hamilton, C. A.

Proc. Applied Superconductivity Conf., IEEE Trans. Magn. MAG-19( 3) : 1291-

1291; May 83.

765
Operation of a Superconducting Anal og-to-Di gi tal Converter at Short Conversion

Times;

Kautz, R. L.

Proc. Applied Superconductivity Conf., IEEE Trans. Magn. MAG-19( 3) : 1186-

1189; May 83.

Superconducting Current Injection Transistor;

VanZeghbroeck, B. J., (Guest Worker)

Appl. Phys. Lett. 42(8) :736-738; Apr 83.

Voltage and Current Expressions for a Two-Junction Superconducting

Interferometer;

Peterson, R. L.; McDonald, D. G.

J. Appl. Phys. 54(2) :992-996; Feb 83.

100 GHz Binary Counter Based on DC SQUIDs;

Hamilton, C. A.

IEEE Electron Device Lett. EDL-3(11) -.335-338; Nov 82.

Electronically Adjustable Delay for Josephson Technology;

Harris, R. E.; Wolf, P.; Moore, D. F.

IEEE Electron Device Lett. EDL-3(9); Sep 82.

High-Speed, Low-Crosstalk Chip Holder for Josephson Integrated Circuits;

Hamilton, C. A.

IEEE Trans. Instrum. Meas. IM-31(2):29; Jun 82.

Book review: Principles of Superconductive Devices and Circuits, by

T. Van Duzer and C. W. Turner;

McDonald, D. G.; Clark, A. F.

Physics Today, p 80; Feb 82.

Analog measurement applications for high speed Josephson switches;

Hamilton, C. A.; Lloyd, Frances L.; Kautz, R. L.

IEEE Trans. Magn. MAG-17(1) :577-82; Jan 82.

A Study of Design Principles for Refrigerators for Low-Power Cryoel ectroni

Devi ces

Zimmerman, J. E.; Sullivan, D. B.

NBS TN 1049; Jan 82.

A Josephson Voltage Standard Using a Series Array of 100 Junctions;

Kautz, R. L.; Costabile, G.

IEEE Trans. Magn. MAG-17, 780; 81.

Magnetic Auditory Evoked Fields: Interhemi spheri c Asymmetry;

Reite, M.; Zimmerman, J. T.; Zimmerman, J. E.

Electroencenphalography and Clinical Neurophysiology 51:388-392; 81.

Superconducting Electronics;

McDonald, D. G.

Phys. Today 34:36-47; 81.

766
Mathematical Modelling of the Impedance of a Josephson Junction Noise

Thermometer;

Peterson, R. L.

J. Appl. Phys., 52(12) :7321-7326; Dec 81.

Design Limitations for Superconducting A/D Converters;

Hamilton, C. A.; Lloyd, F. L. IEEE Trans. Magn. MAG-17(6) :3414-3419; Nov 81.

Chaotic States of rf-Biased Josephson Junctions;

Kautz, R. L.,

J. Appl. Phys. 52(10) :6241-6246; Oct 81.

Modelling the Impedance of a Josephson Junction Noise Thermometer;

Peterson, R. L.; Van Vechten, D. Phys. Review B 24(6) : 3588-3591; Sep 81.

Low Noise Tunnel Junction dc SQUIDs;

Cromar, M. W.; Carelli, P. Appl. Phys. Lett. 38(9) :723-725; May 81.

Measurement of Thermal Properties of Cryocooler Materials;

Zimmerman, J. E.; Sullivan, D. B.; Kautz, R. L.; Hobbs, R. D.

Proc. Conf. on Refrigeration for Cryogenic Sensors and Electronic Systems,

NBS SP 607, p 173-177; May 81.

Operation of a practical SQUID Gradiometer in a Low-power Stirling Cryocooler

Sullivan, D. B.; Zimmerman, J. E.; Ives, J. T.

Conf. on Refrigeration for Cryogenic Sensors and Electronic Systems, NBS SP

607, p 186-194; May 81.

Refrigeration for Cryogenic Sensors and Electronic Systems;

Zimmerman, J. E.; Sullivan, D. B.; McCarthy, S. M., eds.

NBS SP 607; May 81.

The ac Josephson Effect in Hysteric Junctions: Range and Stability

of Phase Lock;

Kautz, R. L.

J. Appl. Phys. 52(5):3528-3541; May 81.

Analog Measurement Applications for High Speed Josephson Switches;

Hamilton, C. A.; Lloyd, F. L.; Kautz, R. L.

IEEE Trans. Magn. MAG 17(1) :577-582; Jan 81.

Behavior of the dc Impedance of an rf-Biased Resistive SQUID;

Van Vechten, D.; Soulen, R. J., Jr.; Peterson, R. L.

Proc. 2nd Int. Conf. on Superconducting Quantum Interference Devices

(SQUIDs), H. D. Wahlbohm; H. Lubbig, eds. New York, Berlin: Walter de

Gruyter & Co., p 569-584; 80.

767
Cryogenics for SQUIDs;

Zimmerman, J. E.

Proc. 2nd Int. Conf. on Superconducting Quantum Interference Devices

(SQUIDs), H. D. Wahlbohm; H. Lubbig, eds. New York, Berlin: Walter de

Gruyter & Co., p 423-443; 80.

Recent Progress in Cryoelectronics;

Sullivan, D. B.; Hamilton, C. A.; Kautz, R. L.

IEEE Trans. Instrum. Meas. IM-29:319; 80.

Space Applications of Superconductivity: Digital Electronics;

Harris, R. E. Cryogenics 20:115; 80.

Transient Pool Boiling of Liquid Helium Using a Temperature Controlled Heater

Surface;

Giarratano, P. J.; Frederick, N. V. Advances in Cryogenic Engineering, K. D. Timmerhaus; H. A. Synder, eds.,

New York: Plenum Press, 25:455-466; 80.

Turn-on Delays in Single Josephson Junction Devices;

Peterson, R. L. Proc. 2nd Int. Conf. on Superconducting Quantum Interference Devices

(SQUIDs), H. D. Wahlbohm; H. Lubbig, eds. New York, Berlin: Walter

de Gruyter & Co., p 685-702; 80.

Picosecond Applications of Josephson Junctions;

McDonald, D. G.; Peterson, R. L.; Hamilton, C. A.; Harris, R. E.;

Kautz, R. L. IEEE Trans. Electron Devices ED-27 ( 10) : 1945-1965 ; Oct 80.

Induced Electronic Currents in the Alaska Oil Pipeline Measured by Gradient

Fluxgate and SQUID Magnetometers;

Campbell, W. H.; Zimmerman, J. E.

IEEE Trans. Geosci . and Remote Sensing GE-18(3) :244-250; Jul 80.

Cryocoolers for Geophysical SQUID Magnetometers;

Zimmerman, J. E.

Proc. SQUID Geophysics Workshop, Los Alamos, NM; Jun 80.

Space Applications of Superconductivity: Instrumentation for Gravitational

and Related Studies;

Peterson, R. L. Cryogenics, p 299-306; Jun 80.

A Superconducting 6-bit Analog-to-Di gital Converter with Operation to 2 x 10 9 Sampl es/Second

Hamilton, C. A.; Lloyd F. L. IEEE Electron Device Lett. EDL-1(5) :92-94; May 80.

Conversion Gain in mm-wave Quasi parti cle Heterodyne Mixers;

Shen, T. M.; Richards, P. L.; Harris, R. E.; Lloyd, F. L.

Appl. Phys. Lett. 36:777-779; May 80.

768
Space Applications of Superconductivity: Microwave and Infrared Detectors;

Hamilton, C. A.,

Cryogenics, p 235-243; May 80.

On a Proposed Josephson-Effeet Voltage Standard at Zero Current Bias;

Kautz, R. L. Applied Phys. Lett. 36(5) :386-388; Mar 80.

Simple-Heati ng-Induced Josephson Effects in Quasi parti cl e-Injected

Superconducting Weak Links;

Kaplan, S. B.

J. Appl. Phys. 51(3) : 1682-1685; Mar 80.

Superconductor Insulator-Superconductor Quasiparticle Junctions as Microwave

Photon Detectors;

Richards, P. L.; Shen, T. M.; Harris, R. E.; Lloyd, F. L.

Appl. Phys. Lett. 36:480-482; Mar 80.

Space Applications of Superconductivity: Low Frequency Superconducting

Sensors;

Zimmerman, J. E.

Cryogenics, p 3-10; Jan 80.

Acoustic Matching of Superconducting Films to Substrates;

Kaplan, S. B.

J. Low Temp. Phys. 37:343-365; 79.

A Milliwatt Stirling Cryocooler for Temperatures below 4 K;

Zimmerman, J. E.; Sullivan, D. B.

Cryogenics 19:170; 79.

Mi nituarization of Normal-State and Superconducting Strip-Lines;

Kautz, R. L. NBS JRES 84:247; 79.

Quasiparticle Heterodyne Mixing in SIS Tunnel Junctions;

Richards, P. L.; Shen, T. M.; Harris, R. E.; Lloyd, F. L.

Appl. Phys. Lett. 34:345; 79.

Superconducting Devices;

Zimmerman, J. E.; Sullivan, D. B.

Yearbook of Science and Technology, McGraw-Hill; 79.

Analysis of Threshold Curves for Superconducting Interferometers;

Peterson, R. L.; Hamilton, C. A.

J. Appl. Phys. 50(12) :8135-8142; Dec 79.

A Superconducting Sampler for Josephson Logic Circuits;

Hamilton, C. A.; Lloyd, F. L.; Peterson, R. L.; Andrews, J. R.

Appl. Phys. Lett. 35:718; Nov 79.

Multiple-Quantum Interference Superconducting Analog-to-digital Converter;

Harris, R. E.; Hamilton, C. A.; Lloyd, F. L.

Appl. Phys. Lett. 35:720; Nov 79.

769
Space Applications of Superconductivity;

Sullivan, D. B.; Vorreiter, J. W.

Cryogenics 19:627-631; Nov 79.

Very Low-power Stirling Cryocoolers Using Plastic and Composite Materials;

Sulllivan, D. B.; Zimmerman, J. E.

Int. J. Refrig. 2:211-213; Nov 79.

Differential Capacitance Sensor as Position Detectors for Magnetic

Suspension Densimeter;

Frederick, N. V.

Rev. Sci. Instrum. 50(9) : 1154; Sep 79.

Analog to Digital Conversion with a SQUID: Conditions for a Countable

Pulse Train;

Peterson, R. L.

J. Appl. Phys. 50:4231; Jun 79.

Cryogenic Refrigeration System;

Zimmerman, James E. PATENT-4 143 520; patented 13 Mar 79.

Quasiparticle Heterodyne Mixing in SIS Tunnel Junctions;

Harris, R. L.; Lloyd, F. L.; Richards, P. L.; Shen, T. M.

Appl. Phys. Lett. 34(5) :345-347; Mar 79.

Sampling Circuit and Method Therefor;

Hami lton, CI ark A.

PAT-APPL-6-020 359; Filed 14 Mar 79.

Attenuation in Superconducting Striplines;

Kautz, R. L. IEEE Trans. Magn. MAG-15:566-569; Jan 79.

High-Speed Superconducting Electronics;

Hamilton, C. A.; Harris, R. E.; Sullivan, D. B.

GOMAC Digest, vol. 7; 78.

Human Magnetic Auditory Evoked Responses;

Reite, M.; Edrich, J.; Zimmerman, J. T.; Zimmerman, J. E.

Electroencephalography and Clinical Neurophysiology 45:114-117; 78.

Magnetic Phenomena of the Central Nervous System;

Reite, Martin; Zimmerman, James

Ann. Rev. Biophys. Bioeng. 7:167-188; 78.

Multiple Magnetic Flux Entry into SQUIDS: A General Way of Examining

the Cos(phi) Conductance;

Peterson, R. L.; Gayley, R. I. Phys. Rev. B 18:1198-1206; Aug 78.

Automatic 300-4 K Temperature Cycling Apparatus;

Hamilton, Clark A.

Rev. Sci. Instrum. Notes 49:674-677; May 78.

770
The Role of Superconductivity in the Space Program: An Assessment of

Present Capabilities and Future Potential;

Sullivan, D. B.

NASA-CR-158701; NBSIR 78-885; May 78.

The Role of Superconductivity in the Space Program: An Assessment of

Present Capabilities and Future Potential;

Sullivan, D. B.

NBSIR 78-885; May 78.

Applications of Closed-Cycle Cryocoolers to Small Superconducting Devices;

Zimmerman, James E.; Flynn, Thomas M.

Proc. Conf. Held at the National Bureau of Standards, Oct 3-4, 1977;

Boulder, CO, NBS SP 508; Apr 78.

RF Instrumentation Based on Superconducting Quantum Interference;

Sullivan, D. B.; Adair, R. T.; Frederi ck, - N. V. Proc. IEEE 66:454-463; Apr 78.

Photolithographic Fabrication of Lead Alloy Josephson Junctions;

Havemann, R. H.; Hamilton, C. A.; Harris, Richard E.

J. Vac. Sci. Technol. 15:392-395; Mar/Apr 78.

Picosecond Pulses on Superconducting Striplines;

Kautz, Richard L. J. Appl. Phys. 49:308-314; Jan 78.

Possible Cryocoolers for SQUID Magnetometers;

Zimmerman, J. E.; Radebaugh, R.; Siegwarth, J. D.

Proc. IC SQUID (1st), Oct 5-8, 1976; Berlin, Germany, Paper R-1044 in

Superconducting Quantum Interference Devices and Their Applications,

p 287-296; 77.

Results, Potentials and Limitations of Josephson Mixer-Receivers at

Millimeter and Long Submi 1 1 imeter Wavelengths;

Edrich, J.; Sullivan, D. B.; McDonald, D. G.

IEEE Trans. Microwave Theory Tech. MTT-25:476; 77.

RF Power Measurements Using Quantum Interference in Superconductors;

Sullivan, D. B.; Frederick, N. V.; Adair, R. T. Proc. IC SQUID (1st), Oct 5-8, 1976; Berlin, Germany, Paper R-1045 in

Superconducting Quantum Interference Devices and Their Applications,

p 355-363; 77.

Superconducting Devices for Metrology and Standards;

Kamper, Robert A.

Chap. 5 in Superconductor Applications: Squids and Machines, New York:

Plenum Press, p 189-247; 77.

Design of a Josephson-Junction Picosecond Pulser;

McDonald, D. G.; Peterson, R. L.; Bender, B. K.

J. 'Appl. Phys. 48:5366-5369; Dec 77.

771
Numerical Evaluation of the Response of a Josephson Tunnel Junction

in an Arbitrary Circuit;

Harris, Richard E.

J. Appl. Phys. 48:5188-5190; Dec 77.

Picosecond Pulse Generator Utilizing a Josephson Junction;

McDonald, Donald G.; Peterson, Robert L.

PAT-APPL-862 311; Filed 20 Dec 77.

A Sampling Circuit and Method Therefor;

Hamilton, Clark A.

PAT-APPL-853 354; Filed 21 Nov 77.

High-Frequency Limitations of the Double-Junction SQUID Amplifier;

Zimmerman, J. E.; Sullivan, D. B.

Appl. Phys. Lett. 31:360-362; Sep 77.

RF Attenuation Measurement System Using a SQUID;

Adair, Robert T.; Frederick, Nolan V.; Sullivan, Donald B.

NBSIR 77-863; Sep 77.

A Low-Noise Josephson Mixer for the 1 mm Wavelength Range;

Edrich, J.; Sullivan, D. B.; McDonald, D. G.

IEEE Trans. Microwave Theory Tech. MTT-25:476-479; Jun 77.

Squid Instruments and Shielding for Low-Level Magnetic Measurements;

Zimmerman, James E.

J. Appl. Phys. 48:702-710; Feb 77.

Advances in the Use of SQUIDS for RF Attenuation Measurement;

Frederick, N. V.; Sullivan, D. B.; Adair, R. T.

IEEE Trans. Magn. MAG-13:361-364; Jan 77.

Analog Computer Studies of Frequency Multiplication and Mixing with

the Josephson Junction;

Risley, A. S.; Johnson, E. G., Jr; Hamilton, C. A.

Proc. 1976 Applied Superconductivity Conf., Stanford, CA,

IEEE Trans. Magn. MAG-13: 381-384; Jan 77.

Can Superconductivity Contribute to the Determination of the Absolute Ampere?;

Sullivan, D. B.; Frederick, N. V. IEEE Trans. Magn. MAG-13:396-399; Jan 77.

Picosecond Pulses from Josephson Junctions: Phenomenological and

Microscopic Analyses;

Peterson, Robert L.; McDonald, Donald G.

IEEE Trans. Magn. MAG-13-.887-890; Jan 77.

Refrigeration for Small Superconducting Devices;

Zimmerman, J. E.; Radebaugh, R.; Siegwarth, J. D.

DKV-annual meeting and joint meeting with the International Institute

of Refrigeration, Munich, Germany, F.R.G, 13 Oct 1976.

Rept No. CONF-7610104-1; 76.

772
The Human Magnetoencephal ogram: Some EEG and Related Correlations;

Reite, Martin; Zimmerman, James E.; Edrich, Jochen; Zimmerman, John

Electroencephalography and Clinical Neurophysiology 40:59-66; 76.

Strong-Coupling Correction to the Jump in the Quasiparticle Current

of a Superconducting Tunnel Junction;

Harris, Richard E.; Dynes, R. C; Ginsberg, D. M.

Phys. Rev. B 14:993-995; Aug 76.

Strong-Coupling Correction to the Low-Frequency Electrical Conductivity of

Superconductors and Josephson Junctions;

Harris, Richard E.; Ginsberg, D. M. ; Dynes, R. C.

Phys. Rev. B 14:990-992; Aug 76.

Intrinsic Response Time of a Josephson Tunnel Junction;

Harris, Richard E. Phys. Rev. B 13:3818-3821; May 76.

RF Applications of the Josephson Effect;

Kamper, Robert A.

Microwave J. 19:39-41; Apr 76.

Modeling Josephson Junctions;

McDonald, D. G.; Johnson, E. G.; Harris, R. E.

Phys. Rev. B 13:1028-1031; Feb 76.

Accurate Rotational Constants, Frequencies, and Wavelengths from

12C 160 Lasers Stabilized by Saturated Absorption;

Petersen, F. R., McDonald, D. G.; Cupp, J. D.; Danielson, B. L.

Proc. Laser Spectroscopy Conf., Jun 25-29, 1973; Vail, CO, p 555-569; 75

Josephson Weak-Link Devices;

Silver, A. H.; Zimmerman, J. E.

Applied Superconductivity, V. L. Newhouse, ed., Academic Press,

p 1-80; 75.

Tests of Cryogenic SQUID for Geomagnetic Field Measurements;

Zimmerman, J. E.; Campbell, W. H. Geophysics 40:269; 75.

Cryogenic Direct Current Comparators and their Applications;

Dziuba, R. F.; Sullivan, D. B.

IEEE Trans. Magn. MAG-11 : 716-719 ; Mar 75.

Phase Slip, Dissipation, Bernoulli Effect, Parametric Capacitance,

and Other Curious Features of the Josephson Effect;

Zimmerman, J. E.

IEEE Trans. Magn. MAG-ll:852-855; Mar 75.

Review of Superconducting Electronics;

Kamper, Robert A.

IEEE Trans. Magn, MAG-11: 141-146; Mar 75.

773
Magnetic Properties of Internally Oxidized Copper;

Fickett, F. R.; Sullivan, D. B.

19th Annual Conf. on Magnetism and Magnetic Materials, 1973; Boston, MA, AIP

Conf. Proc. 18; 74.

Review of Electromagnetic Measurements Using the Josephson Effect;

Kamper, Robert A.

Proc. ISA Int. Instrumentation-Automation Conf., Oct 28-31, 1974; New York,

NY, p 1-8; 74.

RF Attenuation Measurements Using Quantum Interference in Superconductors;

Adair, R. T.; Hoer, C. A.; Kamper, R. A.; Simmonds, M. B.

CPEM Digest, IEE, London, England, p 4-5; 74.

The Relationship of Josephson Junctions to a Unified Standard of

Length and Time;

McDonald, D. 6.; Risley, A. S.; Cupp, J. D.

Proc. 13th Int. Conf. on Low Temperature Physics, Aug 21-25, 1972; Boulder,

CO, Paper in Low Temperature Physics LT 13, p 542-549; 74.

A Low-Temperature Direct-Current Comparator Bridge;

Sullivan, D. B.; Dziuba, R. F.

IEEE Trans. Instrum. Meas. IM-23 : 256-260 ; Dec 74.

RF Attenuation Measurements Using Quantum Interference in

Superconductors;

Adair, R. T.; Simmonds, M. B.; Kamper, R. A.; Hoer, C. A.

IEEE Trans. Instrum. Meas. IM-23 : 375-381 ; Dec 74.

Advances in the Measurement of rf Power and Attenuation Using SQUIDS;

Kamper, R. A.; Simmonds, M. B.; Adair, R. T.; Hoer, C. A.

NBS TN 661; Sep 74.

Josephson Junctions as Radiation Detectors from Mi 11 i hertz to

Terahertz;

McDonald, D. G.

IEEE J. Quantum Electron. QE- 10 : 776-777 ; Sep 74.

An Application of Superconducting Quantum Interference Magnetometers

to Geophysical Prospecting;

Frederick, N. V.; Stanley, W. D.; Zimmerman, J. E.; Dinger, R. J.

IEEE Trans. Geosci. Electron. GE-12: 102-103 ; Jul 74.

Magnetic Studies of Oxidized Impurities in Pure Copper Using a SQUID System;

Fickett, F. R.; Sullivan, D. B.

J. Phys. F 4:900-904; Jun 74.

Josephson Junctions at 45 Times the Energy-Gap Frequency;

McDonald, D. G.; Petersen, F. R.; Cupp, J. D.; Danielson, B. L.;

Johnson, E. G., Jr.

Appl. Phys. Lett. 24:335-337; Apr 74.

774
Low Temperature Direct Current Comparators;

Sullivan, D. B.; Dziuba, R. F.

Rev. Sci. Instrum. 45:517-519; Apr 74.

Superconducting Devices and Materials. Quarterly Literature Survey;

Olien, Neil A.; Goree, William S.; Kamper, Robert A.; Nisenoff, Martin;

Wolf, Stuart A.

NBS Cryogenic Data Center, Boulder, CO, no. 74-1; Jan-Mar 74.

Field-Usable Sharpless Wafers for Josephson Effect Devices at

Millimeter Waves;

Edrich, J.; Cupp, J. D.; McDonald, D. G.

Revue de Physique Appliquee 9:195-197; Jan 74.

Spectral Analysis of a Phase Locked Laser at 891 GHz. An Application

of Josephson Junctions in the Far Infrared;*

Wells, J. S.; McDonald, D. G.; Risley, A. S.; Jarvis, S.; Cupp, J. D.

Revue de Physique Appliques (Supplement to J. de Physique), France,

9:285-292; Jan 74.

Measurement of rf Power and Attenuation Using Superconducting Quantum

Interference Devices;

Kamper, R. A.; Simmonds, M. B.; Hoer, C. A.; Adair, R. T.

NBS TN 643; Aug 73.

Rotational Constants for 12

C 16 0

2 From Beats Between Lamb-DipStabilized Lasers;

Petersen, F. R.; McDonald, D. G.; Cupp, J. D.; Danielson, B. L.

Phys. Rev. Lett. 31:573-576; Aug 73.

Analog-Computer Studies of Mixing and Parametric Effects in Josephson

Juncti ons;

Hamilton, Clark A.

J. Appl. Phys. 44:2371-2377; May 73.

Portable Helium Dewars for Use with Superconducting Magnetometers;

Zimmerman, J. E.; Siegwarth, J. D.

Cryogenics 13:158-159; Mar 73.

Superconducting Devices and Materials;

Goree, William S.; Nisenoff, Martin; Wolf, Stuart A.; Kamper, Robert A.

Olien, Neil A.

NBS Cryogenic Data Center; Boulder, CO, Quarterly literature survey,

no. 73-1; Jan-Mar 73.

A New Technique for RF Measurements Using Superconductors;

Kamper, R. A.; Simmonds, M. B.; Adair, R. T.; Hoer, C. A.

Proc. IEEE 61:121-122; Jan 73.

Flexible Laminates for Thermally Grounded Terminal Strips and Shielded

Electrical Leads at Low Temperatures;

Radebaugh, Ray; Frederick, N. V.; Siegwarth, J. D.

Cryogenics 13:41-43; Jan 73.

775
Possible Parametric Capacitance in Josephson Junctions;

Zimmerman, J. E.

Phys. Lett. 42A:375-376; Jan 73.

A Review of the Properties and Applications of Superconducting Point

Contacts;

Zimmerman, J. E.

Proc. Conf. on Applied Superconductivity, May 1-3, 1972; Annapolis, MD,

p 544-561; 72.

Precise Electrical Measurements at Low Temperature;

Sullivan, D. B.

Proc. Applied Superconductivity Conf., May 1-3, 1972; Annapolis, MD;

p 631-639; 72.

Quantum Mechanical Measurement of rf Attenuation;

Kamper, R. A.; Simmonds, M. B.; Adair, R. T.; Hoer, C. A.

Proc. 1972 Applied Superconductivity Conf., May 1-3, 1972; Annapolis, MD,

p 696-700; 72.

Computation of Spectral Data for a Josephson Junction Circuit;

Johnson, E. 6., Jr; McDonald, D. G.

NBS TN 627; Nov 72.

Developments in Cryoelectronics;

Kamper, Robert A.; Sullivan, Donald B.

NBS TN 630; Nov 72.

Superconducting Quantum Interference Devices: an Operational Guide

for rf-Biased Systems;

Sullivan, D. B.

NBS TN 629; Nov 72.

Analog Computer Studies of Subharmonic Steps in Superconducting Weak

Li nks;

Hamilton, C. A.; Johnson, E. G., Jr.

Phys. Lett. A 41:393-394; Oct 72.

Broadband Superconducting Quantum Magnetometer;

Kamper, R. A.; Simmonds, M. B.

Appl. Phys. Lett. 20:270-272; Apr 72.

Four-Hundredth-Order Harmonic Mixing of Microwave and Infrared Laser

Radiation Using a Josephson Junction and a Maser;

McDonald, D. G.; Risley, A. S.; Cupp, J. D.; Evenson, K. M.;

Ashley, J. R.

Appl. Phys. Lett. 20:296-299; Apr 72.

Low Temperature Voltage Divider and Null Detector;

Sullivan, D. B.

Rev. Sci. Instrum. 43:499-505; Mar 72.

776
Superconducting Devices and Materials. A Literature Survey;

Goree, W. S.; Kamper, R. A.; Olien, Neil A.

NBS Cryogenic Data Center; Boulder, CO, Quarterly reports; 4 issues; Mar 72.

Josephson Effect Devices and Low-Frequency Field Sensing;

Zimmerman, J. E.

Cryogenics 12:19-31; Feb 72.

A Mechanical Superconducting Switch for Low Temperature

Instrumental on; Siegwarth, J. D.; Sullivan, D. B.

Rev. Sci. Instrum. 43:153-154; Jan 72.

Superconducting Materials;

Kamper, R. A.

Electronics Design Materials, p 71-79; 71.

Survey of Noise Thermometry;

Kamper, Robert A.

Proc. 5th Symposium on Temperature, Its Measurement and Control in Science

and Industry, Jun 21-24, 1971; Washington, DC; 71.

Mechanical Analogs of Time Dependent Josephson Phenomena;

Sullivan, D. B.; Zimmerman, J. E. American J. Phys. 39:1504-1517; Dec 71.

Sensitivity Enhancement of Superconducting Quantum Interference

Devices through the Use of Fracti onal -Turn Loops;

Zimmerman, J. E. J. Appl. Phys. 24:4483-4487; Oct 71.

Observation of Noise Temperature in the Millikelvin Range;

Kamper, R. A.; Siegwarth, J. D.; Radebaugh, R.; Zimmerman, J. E.

Proc. IEEE 59:1368-1369; Sep 71.

Miniature Ultrasensitive Superconducting Magnetic Gradiometer and

Its Use in Cardiography and Other Applications;

Zimmerman, J. E.; Frederick, N. V. Appl. Phys. Lett. 19:16-19; Jul 71.

Resistance of a Silicon Bronze at Low Temperatures;

Sullivan, D. B.

Rev. Sci. Instrum. 42:612-613; May 71.

Harmonic Mixing of Microwave and Far-Infrared Laser Radiation using

a Josephson Junction;

McDonald, D. G.; Risley, A. S.; Cupp, J. D.; Evenson, K. M.

Appl. Phys. Lett. 18:162-164; Feb 71.

Hi gh-Frequency Limit of the Josephson Effect;

McDonald, D. G.; Evenson, K. M.; Wells, J. S.; Cupp, J. D.

J. Appl. Phys. 42:179-181; Jan 71.

777
Noise Thermometry with the Josephson Effect;

Kamper, R. A.; Zimmerman, J. E.

J. Appl. Phys. 42:132-136; Jan 71.

Recent Developments in Superconducting Devices;

Zimmerman, J. E.

J. Appl. Phys. 42:30-37; Jan 71.

Cryoelectronics;

Kamper, R. A.

Proc. Helium Society Symposium, Mar 23-24, 1970; Washington, DC, p 68-82;

70.

Generation of Harmonics and Subharmonics of the Josephson

Oscillation;

Sullivan, D. B.; Peterson, Robert L.; Kose, V. E.; Zimmerman, J. E.

J. Appl. Phys. 41:4865-4873; Nov 70.

Magnetocardi ograms Taken Inside A Shielded Room With A Superconducting

Point-Contact Magnetometer;

Cohen, David; Edelsack, Edgar A.; Zimmerman, James E.

Appl. Phys. Lett. 16:278-280; Apr 70.

Some Applications Of The Josephson Effect;

Kamper, R. A.; Mullen, L. 0.; Sullivan, D. B.

NASA-CR-1565; NBS TN 381; Mar 70.

Influence of External Noise On Microwave-Induced Josephson Steps;

Kose, V. E.; Sullivan, D. B.

J. Appl. Phys. 41:169-174; Jan 70.

The Josephson Effect;

Kamper, Robert A.

IEEE Trans. Electron Devices ED-16:840-844 ; Oct 69.

Harmonic Generation and Submi 1 1 imeter Wave Mixing With The Josephson

Effect;

McDonald, D. G.; Kose, V. E.; Evenson, K. M.; Wells, J. S.; Cupp, J. D.

Appl. Phys. Lett. 15:121-122; 15 Aug 69.

Fabrication Of Tunnel Junctions On Niobium Films;

Mullen, L. 0.; Sullivan, D. B.

J. Appl. Phys. 40:2115-2117; Apr 69.

Cryoelectronics;

Kamper, R. A.

Cryogenics, 9:20-25; Feb 69.

Contribution of Thermal Noise to the Line-Width of Josephson

Radiation from Superconducting Point Contacts;

Silver, A. H.; Zimmerman, J. E.; Kamper, R. A.

Appl. Phys. Lett. 11:209-211; 15 Sep 67.

778
Mi 1 1 i degree Noise Thermometry;

Kamper, R. A.

Proc. Symposium on the Physics of Superconducting Devices (ONR report),

Charlottesville, VA; Apr 67.

779
SUPERCONDUCTORS AND MAGNETIC MATERIALS

Magnetic Field Effects on Tensile Behavior of Alloys 304 and 310 at 4 K

Reed, R. P.; Arvidson, J. M.; Ekin, J. W.; Schoon, R. H. Proc. Int. Cryogenic Materials Conf.; May 11-14, 1982; Kobe Japan; p 33-36;

83.

Conductors for Advanced Energy Systems Annual Report 1982;

Ficket, F. R. INCRA Research Report, Paper#321A; Int. Copper Research Assoc., 708 3rd

Ave., NY 10017; p 1-97; Aug 83.

Electrical Properties;

Fickett, F. R. Chapt. 5 in Materials at Low Temperatures, R. Reed; A. F. Clark, eds.,

American Society for Metals, Metals Park, OH 44073; p 163-201; Jun 83.

Magnetic Properties;

Fickett, F. R.; Goldfarb, R. B.

Chap. 6 in Materials at Low Temperatures, R. Reed; A. F. Clark, eds.,

American Society of Metals, Metals Park, OH 44073; p 203-235; Jun 83.

Superconductors;

Ekin, J. W.

Chapt. 13 in Materials at Low Temperatures, R. Reed; A. F. Clark, eds.,

American Society of Metals, Metals Park, OH 44073, p 465-513; Jun 83.

Thermal Expansion;

Clark, A. F.

Chapt. 3 in Materials at Low Temperatures, R. Reed; A. F. Clark, eds.,

American Society for Metals, Metals Park, OH 44073, p 75-132; Jun 83.

Effect of Stainless Steel Reinforcement on the Critical Current Versus Strain

Characteristic of Mul tifilamentary Nb

3

Sn Superconductors;

Ekin, J. W.

J. Appl. Phys. 54(5) :2869-2871; May 83.

J-B-T-e Interaction in A15, Bl, and C15 Crystal Structure Superconductors;

Ekin, J. W.

Proc. Applied Superconductivity Conf., IEEE Trans. Magn. MAG-19: 900-902;

May 83.

Mul ti f i 1 amentary Nb-Nbo Composite by Liquid Infiltration Method:

Superconducting, Metallurgical, and Mechanical Properties;

Hong, M.; Hull, G. W., Jr.; Holthuis, J. T.; Hazzenzahl , W. V.; Ekin, J. W.

Proc. Applied Superconductivity Conf., IEEE Trans. Magn. MAG-19(3) :912-916

May 83.

Oxygen-Free Copper at 4K;

Fickett, F. R. Proc. Applied Superconductivity Conf., IEEE Trans. Magn. MAG-19: 228-231;

May 83.

780
Properties of NbN Films Crystallized from the Amorphous State;

Gavaler, J. R. ; Greggi, J.; Wilmer, R. ; Ekin, J. W.

Proc. Applied Superconductivity Conf., IEEE Trans. Magn. MAG-19(3) :418-421;

May 83.

The Effect of Field Orientation on Current Transfer in Mul ti filamentary

Superconductors;

Goodrich, L. F.

Proc. Applied Superconductivity Conf., IEEE Trans. Magn. MAG-19(3) :244-247;

May 83.

Four-Dimensional J-B-T-e Critical Surface for Superconductivity;

Ekin, J. W.

J. Appl . Phys. 54(1) :303-306; Jan 83.

Proc. International Cryogenic Materials Conference; May 11-14, 1982; Kobe,

Japan;

Clark, A. F.; Tachikawa, K., eds.

Butterworths; Jan 83.

Effect of Strain on the Critical Current and Critical Field of Bl Structure

NbN Superconductors;

Ekin, J. W.

Appl. Phys. Lett. 41(10) :996; Nov 82.

Effect of Strain on the Critical Current of Sputtered NbN Films;

Ekin, J. W.; Gavaler, J. R.; Greggi, J.

Bull. Am. Phys. Soc. p 8-12; Mar 1982 and Appl. Phys. Lett. 41(10) :996-998;

Nov 82.

Spin-Freezing below the Ferromagnetic Transition Determined by the Imaginary

Component of ac Magnetic Susceptibility

Goldfarb, R. B.; Fickett, F. R.; Rao, K.V.; Chen, H. S.

J. Appl. Phys. 53(11) :7687; Nov 82.

Development of Standards for Superconductors;

Clark, A. F.

NBSIR 82-1678; Jul 82.

Effect of Twist Pitch on Short-Sample V-I Characteristics of Mul tifilamentary

Superconductors;

Goodrich, L. F.; Ekin, J. W.; Fickett, F. R.

Adv. Cryo. Eng. 28:571-58; New York: Plenum Press; Jul 82.

Low Temperature Material Perspective;

Fickett, F. R. Adv. Cryo. Eng. 28:1-16; Jul 82.

Training Studies of Epoxy-Impregnated Superconductor Windings;

Ekin, J. W; Pittman, E. S.; Supercynski , M. J.; Waltman, D. J.

Adv. Cryo, Eng., 28:719-728, New York: Plenum Press; Jul 82.

781
Critical Current Measurement: A Compendium of Experimental Effects;

Goodrich, L. F.; Fickett, F. F.

Cryogenics, p 225-242; May 82.

Effect of Strain on the Critical Parameters of V2(Hf,Zr) Laves Phase Composite

Superconductors;

Ekin, J. W.

Appl. Phys. Lett. 40(9) :844-846; May 82.

Electrical Properties of Materials and Their Measurement at Low Temperatures;

Fickett, F. R. NBS TN 1053; Mar 82.

Electric and Magnetic Properties of CuSn and CuNi Alloys at 4 K; Fickett, F. R.

Cryogenics 22:135; Mar 82.

Further Evidence for a Spin-Glass Phase Transition in Amorphous Fe-Mn-P-B-Al

Alloys;

Goldfarb, R. B.; Rao, K. V.; Chen, H. S.; Patton, C. E. J. Appl . Phys. 53:2217; Mar 82.

Electrical and Magnetic Properties of Internally Oxidized Copper and Dilute

Copper- Iron Alloys;

Fickett, F. R.

J. Phys. F: Met. Phys. 12:1953-1969; Jan 82.

Effect of Strain on the Critical Current of Nb-Hf/Cu-Sn-Ga Multi filamentary

Superconductors;

Ekin, J. W.; Sekine, H.; Tachikawa, K.

J. Appl. Phys. 52:6252; 81.

Lap Joint Resistance and Intrinsic Critical Current Measurements on a NbTi

Superconducting Wire;

Goodrich, L. F.; Ekin, J. W.

IEEE Trans. Magn. MAG-17:69; 81.

Magnetic Susceptibility Studies of Amorphous Ni-Mn-P-B-Al Alloys;

Goldfarb, R. B.; Rao, K. V.; Fickett, F. R.; Chen, H. S.

J. Appl. Phys. 52:1744; 81.

Mechanical Properties and Strain Effects in Superconductors;

Ekin, J. W.

Chap. 7 in Superconducting Materials Science, S. Foner and B. Schwartz,

eds., New York: Plenum Press, p 455-509; 81.

Miniature Multi pin Electrical Feedthrough for Vacuum Use;

Goldfarb, R. B.

Cryogenics 21:746; 81.

Strain Scaling Law for Flux Pining in NbTi, Nb Sn, Nb-Hf/Cu-Sn-Ga, V Ga, and

Nb Ge;

3 3 Ekin, J. W.

IEEE Trans. Magn. MAG-17:658; 81.

782
Structural Materials for Large Superconducting Magnets;

Pickett, F. R. ; McHenry, H. I. IEEE Trans. Magn. MAG-17:2297; 81.

Superparamagneti sm and Spin-Glass Freezing in Nickel -Manganese Alloys;

Goldfarb, R. B.; Patton, C. E.

Phys. Rev. B 24:1360; 81.

NBS Superconductor Standardization Program;

Fickett, F. R.; Goodrich, L. F.

Proc. 1980 Superconducting MHD Magnet Design Conf. (Francis Bitter National

Magnet Laboratory, MIT); Oct 81.

Thermal Expansion of Mul tifilamentary Nb 3 Sn and V

3Ga Superconductive Cables

and Fiberglass-Epoxy and Cotton-Phenolic Composite Materials;

Fujii, G.; Ranney, M. A.; Clark, A. F.

Jap. J. Appl . Phys. 20, L267-L270; Apr 81.

Thermal Expansion of Several Materials for Superconducting Magnets;

Clark, A. F.; Fujii, G.; Ranney, M. A.

Proc. 7th Magnet Technology Conf., Karlsruhe, Germany, IEEE Trans. Magn.

MAG-17:2316; Apr 81.

Advances in Cryogenic Engineering - Materials, Vol. 26;

Clark, A. F.; Reed, R. P., eds.

Proc. 3rd Int. Cryogenic Mater. Conf., Aug 1979; Madison, WI, New York:

Plenum Press; 80.

Definitions of Terms for Practical Superconductors,

4. Josephson Phenomena;

Fickett, F. R.; Kaplan, S. B.; Powell, R. L.; Radebaugh, R.; Clark, A. F.

Cryogenics 20:319-325; 80.

Development of Standards for Superconductors;

Fickett, F. R.; Clark, A. F.

Proc. 8th Int. Cryogenic Eng. Conf., Jun 1980; Genova; IPC Science and

Technology Press, Guildford, England, p 494-8; 80.

Filamentary A15 Superconductors;

Suenaga, M.; Clark, A. F., eds.

New York: Plenum Press; 80.

Processing Limits for Ultrafine Multifilament Nb Sn;

Ho, J. C; Oberly, C. E.; Garrett, H. J.; Walker, M. S.; Zeitlin, B. A.;

Ekin, J. W.

Proc. Int. Cryo. Mater. Cong., 1979; Madison, WI, Adv. Cryogenic. Eng.

26:358; 80.

Strain Scaling Law and the Prediction of Unixial and Bending Strain Effects;

Ekin, J. W.

Filamentary A15 Superconductors, M. Suenaga and A. F. Clark, eds., New York

Plenum Press; p 455-509; 80.

783
Strain Scaling Law for Flux Pinning in Practical Superconductors. Part I: Basic Relationship and Application to h^Sn Conductors;

Ekin, J. W.

Cryogenics 20:611; 80.

Tensile, Fracture Toughness and Magnetization Testing in Cast 316-L Stainless

Steel and Its Weldment;

Genens, L.; Kim, S. H.; Wang, S. T.; Reed, R. P., Fickett, F. R. Proc. 8th Int. Cryo. Eng. Conf. (IPC Science and Technology Press, paper 3E)

20 p; 80.

Thermal Expansion of Cryogenic-Grade Glass-Epoxy Laminates; Materials Studies

of Magnetic Fusion Energy Applications at Low Temperatures - III;

Ranney, M. A.; Clark, A. F.

NBSIR 80-1627:405; 80.

Training of Epoxy- Impregnated Superconductor Windings;

Ekin, J. W.; Schramm, R. E.; Superczynski , M. J.

Proc. 3rd Int. Cryogenic Mater. Conf., Aug 1979; Madison, WI; Adv. Cryo.

Eng. 26:677; 80.

Development of Standards for Superconductors;

Fickett, F. R.; Goodrich, L. F.; Clark, A. F.

NBSIR 80-1642; Dec 80.

Effect of Thermal Contraction of Sample Holder Material on Critical

Current Measurement;

Fujii, G.; Ekin, J. W.; Radebaugh, R.; Clark, A. F.

Advances in Cryogenic Engineering - Materials, Vol. 26, New York: Plenum

Press; p 589-98; 80. Also published as Technical Report A-1074, Institute

of Solid State Physics, Univ. of Tokyo; Aug 80.

Development of Standards for Practical Superconductors;

Clark, A. F.

Proc. Superconductivity Technical Exchange, PIC-E:E-SC 209/1, Naval Research

Laboratory, Washington, DC; p 103-14; Apr 80.

A Convenient Standard for Low-Field Susceptibility Calibration;

Rosenblum, J.; Larson, E.; Hoblitt, R.; Fickett, F. R. Rev. Sci. Instrum 50:1027; 79.

Effect of Strain on Critical Current of Nb Ge;

Ekin, J. W.; Braginski, A. I. IEEE Trans. Magn. MAG-15 :509; 79.

Effect of Strain on Epoxy-Impregnated Superconducting Composites;

Ekin, J. W.; Schramm, R. E.; Clark, A. F.

Nonmetallic Materials and Composites at Low Temperatures; A. F. Clark;

R. P. Reed; G. Hartwig, eds., New York: Plenum Press, p 301-8; 79.

Materials for Superconducting Magnet Systems;

Fickett, F. R.; Reed, R. P., eds.

Traverse City, MI: Bel four-Stul en, Inc.; 79.

784
Nonmetal 1 ic Materials and Composites at Low Temperatures;

Clark, A. F. ; Reed, R. P.; Hartwig, G. , eds.

Proc. ICMC Symposium, Jul 1978; Munich, Germany, New York: Plenum Press; 79.

Space Applications of Superconductivity: High Field Magnets;

Fickett, F. R.

Cryogenics 19:691-701; 79.

Standards for Superconductors;

Fickett, F. R.; Clark, A. F.,

DoE Conf. 79-0854, Proc. Mechanical and Magnetic Energy Storage, U.S. Dept.

of Energy, Washington, DC, p 3-8; 79.

Strain Dependence of the Critical Current and Critical Field in

Multi filamentary Nb Sn Composites;

Ekin, J. W.

3 IEEE Trans. Magn. MAG-15:197; 79.

Structures, Insulators, and Conductors for Large Superconducting Magnets;

Fickett, F. R.; Reed, R. P.; Dalder, E. N. C.

J. Nucl. Mat. 85 and 86:353-60; 79.

The Development of Standards for Practical Superconductors

Clark, A. F.; Ekin, J. W.; Radebaugh, R.; Read, D. T.

IEEE Trans. Magn. MAG-15:224-7; 79.

Development of Standards for Superconductors;

Fickett, F. R.; Clark, A. F.

NBSIR 80-1629; Dec 79.

Material Studies for Superconducting Machinery Coil Composites;

Ekin, J. W.; Kasen, M. B.; Read, D. T; Schramm, R. E.; Tobler, R. L.;

Clark, A. F.

NBSIR 80-1633; Nov 79.

A Standards Program for AC Losses in Superconductors;

Radebaugh, R.; Fujii, G.; Read, D. T.; Clark, A. F.

Int. Congress of Refrigeration, I I R Al/2-10:l-4; Sep 79.

Definitions of Terms for Practical Superconductors.

3. Fabrication, Stabilization, and Transient Losses;

Read, D. T.; Ekin, J. W.; Powell, R. L.; Clark, A. F.

Crogenics 19(6) :327-32; Jun 79.

Magnetic Properties of the 'Nonmagnetic' Stainless Steels;

Fickett, F. R.

NBSIR 79-1609; Jun 79.

Materials for Superconducting Magnets for MHD Power Systems, a Usage Survey

and a Proposed Research Program;

Reed, R. P.; McHenry, H. I.; Kasen, M. B.; Fickett, F. R.; Dalder, E. N. C.

MIT Program Report on MHD; Jun 79.

785
Materials Studies for Magnetic Fusion Energy Applications at Low

Temperatures - II; Fickett, F. R. , ed.

NBSIR 79-1609; Jun 79.

Review of the 1978 NBS/DoE Workshop on Materials at Low Temperatures;

Fickett, F. R.; Reed, R. P. Proc. 1st Topical Meeting on Fusion Reactor Materials, Miami Beach, FL,

p 352; Jan 79.

Advances in Cryogenic Engineering, Vol. 24;

Timmerhaus, K. D.; Reed, R. P.; Clark, A. F., eds.

Proc. 2nd Int. Cryogenic Mater. Conf., Aug 1977; Boulder, CO; New York:

Plenum Press; 78.

Current Transfer in Mul ti filamentary Superconductors. I. Theory;

Ekin, J. W.

J. Appl. Phys. 49(6) :3406-l; 78.

Effects of Stress on Practical Superconductors;

Clark, A. F.

MT-6, Proc. Int. Conf. on Magn. Tech., Alpha, Bratislava, Czechoslovakia,

p 612-18; 78.

Fatigue and Stress Effects in NbTi and Nb

3

Sn Mul ti filamentary Superconductors;

Ekin, J. W.

Proc. Int. Cryogenic Mater. Conf., 1977; Boulder, CO; Adv. in Cry. Eng.

24:306; 78.

Low Temperature Specific Heat of Two Stainless Steels;

Ho, J. C; King, G. B.; Fickett, F. R. Cryogenics 18:296; 78.

Properties of a Superconducting Coil Composite and Its Components;

Clark, A. F.; Arp, V. D.; Ekin, J. W.

MT-6, Proc. Int. Conf. on Magn. Tech., Alpha, Bratislava, Czechoslovakia,

p 673-79; 78.

Special Purpose Materials: An Assessment of Needs and the Role of these

Materials in the National Program;

Gold, R. E. ; Fickett, F. R.; et al

Proc. 3rd Topical Meeting on the Technology of Controlled Nuclear Fusion,

DoE Conf-780508; 78.

Magnet Materials for Fusion Energy Section of: "The Fusion Reactor Materials

Program Plan, Section IV, Special Purpose Materials";

Fickett F. R.

U.S. DoE Report DoE/ET-0032/4; Jul 78.

Current Transfer in Mul ti filamentary Superconductors.

II. Experimental Results;

Ekin, J. W.; Clark, A. F.; Ho, J. C.

J. Appl. Phys. 49(6) :3410-1; Jun 78.

786
High Field Magnets;

Fickett, F. R. Chap. 2 in The Role of Superconductivity in the Space Program: An

Assessment of Present Capabilities and Future Potential, NBSIR 78-885;

May 78.

Materials Studies for Magnetic Fusion Energy Applications at Low

Temperatures - I; Fickett, F. R. ; Reed, R. P. NBSIR 78-884; Apr 78.

Definitions of Terms for Practical Superconductors. 2. Critical Parameters

Powell , R. L.; Clark, A. F.

Cryogenics 18(3) : 137-41 ; Mar 78.

Investigation of a Practical Superconductor with a Copper Matrix;

Fickett, F. R.

I NCRA Annual Report, Project No. 255; Jan 78.

A Low Temperature Materials Research Program for Magnetic Fusion Energy;

Fickett, F. R.; Kasen, M. B.; McHenry, H. I.; Reed, R. P. Adv. Cryogenic Eng. 24:52; 77.

A Review of the NBS-ERDA Workshop on Materials at Low Temperatures;

Fickett, F. R. ; Reed, R. P.

Proc. 7th Symposium on Engineering Problems of Fusion Research, IEEE Pub.

No. 77CH1267-4-NPS:1506-1509; 77.

A Simple Method for Producing High Conductivity Copper for Low Temperature

Appl i cations;

Rosenblum, S.; Steyert, W. A.; Fickett, F. R. Cryogenics 17:645; 77.

Mechanisms for Critical -Current Degradation in NbTi and Nb3Sn and NbTi

Mul ti f i 1 amentary Wi res

;

Ekin, J. W.

Proc. 1976 Applied Superconductivity Conf., IEEE Trans. Magn. MAG-13:127;

77.

Studies of Superconducting Wires from Niobium Precipitated in Copper-TinNiobium Alloys;

Fickett, F. R.; Sparks, L. L.; Kasen, M. G.

Manufacture of Superconducting Materials, R. W. Meyerhoff, ed., American

Society for Metals, Metals Park, OH, p 164; 77.

The Low Temperature Tensile Behavior of Copper-Stabilized Niobium-Titanium

Superconducting Wire;

Reed, R. P.; Mikesell, R. P.; Clark, A. F.

Advances in Cryogenic Engineering, vol. 22, New York: Plenum Press,

p 463-471; 77.

787
Definition of Terms for Practical Superconductors. 1. Fundamental States and

Flux Phenomena;

Powell , R. L. ; Clark, A. F.

Cryogenics 17(12) :697-701; Dec 77.

Magnetic Fusion Energy Low Temperature Materials Program - A Survey;

Reed, R. P.; Fickett, F. R.; Kasen, M. B.; McHenry, H. I. Report to ERDA Division of Magnetic Fusion Energy; Mar 77.

Defining Critical Current;

Clark, A. F.; Ekin, J. W.

IEEE Trans. Magn. MAG-13U ) :38-40; Jan 77.

Advances in Cryogenic Engineering - Vol. 22;

Timmerhaus, K. D.; Reed, R. P.; Clark, A. F., eds.

Proc. 1st Int. Cryogenic Mater. Conf., Aug. 1975; Kingston, Ontario,

New York: Plenum Press; 76.

A Research Program on the Properties of Structural Materials at 4 K; Reed, R. P.; Clark, A. F.; van Reuth, E. C.

Adv. Cryo. Eng., New York: Plenum Press, 22:1-8; 76.

Effect of Strain on the Critical Current of Nb3Sn and NbTi Multi filamentary

Composi te Wi res

Ekin, J. W.; Clark, A. F.

AIP Conf. Proc. no. 34, New York: Amer. Inst. Phys., p 81-83; 76.

Effect of Stress on the Critical Current of Nb

3

Sn Multi filamentary Composite

Wi re; Ekin, J. W.

Appl . Phys. Lett. 29:216; 76.

Effect of Stress on the Critical Current of NbTi Multi filamnetary

Composite Wire;

Ekin, J. W.; Fickett, F. R.; Clark, A. F.

Adv. Cryo. Eng., New York: Plenum Press, 22:449-452; 76.

Effect of Stress on the Critical Current of NbTi Mul ti filamentary

Composite Wire;

Ekin, J. W.; Fickett, F. R.; Clark, A. F.

Proc. Int. Cryogenic Mater. Conf., 1975; Kingston, Ontario; Adv. in

Cryogenic Eng., New York: Plenum Press 22:449; 76.

Magnetic and Electrical Properties of Internally Oxidized FeCu Alloys;

Fickett, F. R.

Amer. Inst. Phys. Conf. Proc. 34:25; 76.

On Lysozyme as a Possible High-Temperature Superconductor;

Sorenson, C. M.; Fickett, F. R.; Mockler, R. C; O'Sullivan, W. J.;

Scott, J. F.

J. Phys. C: Solid State Phys. 9:L251; 76.

788
Properties of Nonsuperconducting Technical Solids at Low Temperatures - An

Update;

Fickett, F. R. Proc. 5th Int. Conf. on Magn. Tech. (MT-5), Rome, Laboratori Nazionali del

CNEN, Frascati, Italy, p 659; 76.

Structural Materials for Cryogenic Appi cations;

Fickett, F. R. Proc. 6th Int. Cryogenic Eng. Conf., May 11-14, 1976; Grenoble, France; IPC

Science and Technology Press, p 20; 76.

Stress Effects in Superconductors;

Clark, A. F.

Cryogenics 16(10) :632-3; Oct 76.

Controlled Thermonuclear Reactors: A Prospective Large- Scale Use of Pure

Copper;

Fickett, F. R.

INCRA Research Report; Aug 76.

Characterization of a Superconducting Coil Composite and Its Components;

Clark, A. F.; Weston, W. F.; Arp, V. D.; Hust, J. G.; Trapani , R. J.

NBSIR 76-837; Jul 76.

Low Temperature Thermal Expansion of Barium Ferrite;

Clark, A. F.; Haynes, W. M.; Deason, V. A.; Trapani, R. J.

Cryogenics 16(3) :267-270; May 76.

A Preliminary Investigation of the Behavior of High Purity Copper in High

Magnetic Fields and a Final Summary of Project 186;

Fickett, F. R. INCRA Annual Report, Project No. 186C; Mar 76.

A Technique for Preparing Homogeneous Bulk Samples of Concentrated Alloys;

Ekin, J. W.; Deason, V. A.

Rev. Sci . Instr. 46:327; 75.

Critical Currents in Granular Superconductors;

Ekin, J. W.

Phys. Rev. B 12:2676; 75.

The Magnetic Coupling Force of the Superconducting dc Transformer;

Ekin, J. W.; Clem, J. R. Phys. Rev. B 12:1753; 75.

Materials Research for Superconducting Machinery - IV; Semi -Annual Tech.

Rept., Mar-Sep. 1975;

Reed, R. P.; Clark, A. F.; vanReuth, E. C, eds.

Advanced Research Projects Agency, Arlington, VA ADA019230; Oct 75.

Materials Research for Superconducting Machinery - III; Semi-Annual Tech.

Rept., Sep. 74-Mar. 75;

Reed, R. P.; Clark, A. F.; van Reuth, E. C, eds.

Advanced Research Projects Agency, Arlington, VA, ADA012365; Apr 75.

789
Magnetic Properties of Internally Oxidized Copper;

Fickett, F. R., Sullivan, D. B.

AIP Conf. Proc. 18:740; 74.

Magnetic Studies of Oxidized Impurities in Pure Copper Using a SQUID System;

Fickett, F. R.; Sullivan, D. B.

J. Phys. F 4:900; 74.

Magnetothermal Conductivity;

Fickett, F. R.; Sparks, L. L.

NBSIR 74-393; 74.

Magnetothermal Conductivity;

Sparks, L. L; Fickett, F.R.

NBSIR 74-359; 74.

Oxygen Annealing of Copper: A Review;

Fickett, F. R.

Mat. Sci. and Eng. 14:199; 74.

U. S. Programs on Large Scale Applications of Superconductivity;

Powell, R. L.; Fickett, F. R.; Birmingham, B. W.

Chap. 17 in Superconducting Machines and Devices -- Large Systems

Applications; Proc. NATO Advanced Study Institute, Sep 5-14, 1973; Entreves,

Italy; S. Foner and B. B. Schwartz, eds., New York: Plenum Press, p 651-675;

74.

Materials Research for Superconducting Machinery - II, Semi-Annual Tech.

Rept., Mar. -Sep. 74;

Clark, A. F.; Reed, R. P.; van Reuth, E. C, eds.

Advanced Research Projects Agency, Arlington, VA, ADA004586; Oct 74.

A Preliminary Investigation of the Behavior of High Purity Copper in High

Magnetic Fields;

Fickett, F. R.

INCRA Annual Report, Project No. 186B; Aug 74.

Materials Research for Superconducting Machinery, Semi -Annual Tech. Rept., Sep

1973-Mar 1974;

Clark, A. F.; Reed, R. P.; van Reuth, E. C, eds.

Advanced Research Projects Agency, Arlington, VA, AD780596; Mar 74.

Characterization of a Superconducting Coil Composite;

Fowlkes, C. W.; Angerhofer, P. E.; Newton, R. N,; Clark, A. F.

NBSIR 73-349; Dec 73.

Superconducting Levitation of High Speed Vehicles;

Arp, V. D.; Clark, A. F.; Flynn, T. M.

Transport. Eng. J., ASCE 99:873-85; Nov 73.

A Compilation and Evaluation of Mechanical, Thermal and Electrical Properties

of Selected Polymers;

Schramm, R. E. ; Clark, A. F.; Reed, R. P. NBS MN 132, Washington, DC: U.S. GPO; Sep 73.

790
A Preliminary Investigation of the Behavior of High Purity Copper in High

Mangetic Fields;

Fickett, F. R.

INCRA Annual Report, Project No. 186A; Aug 73.

Mechanical, Thermal, and Electrical Properties of Selected Polymers;

Reed, R. P.; Schramm, R. Clark, A. F.

Cryogenics 13:67-82; Feb 73.

Some Applications of Cryogenics to High Speed Ground Transportation;

Arp, V. D.; Clark, A. F.; Flynn, T. M.

NBS TN 635; Feb 73.

Characterization of High Purity Metals by the Eddy Current Decay Method;

Clark, A. F.; Deason, V. A.; Powell, R. L. Cryogenics 12:35; 72.

Magnetoresistivity of Copper and Aluminum at Cryogenic Temperatures;

Fickett, F. R. Proc. 4th Int. Conf. on Magnet Technology, Sep 72, AEC C0NF-720908:498; 72.

Material Variability as Measured by Low Temperature Electrical Resistivity;

Clark, A. F.; Tryon, P. V. Cryogenics 12:451-61; Dec 72.

Properties of Nonsuperconducting Technical Solids at Low Temperatures;

Fickett, F. R. Proc. 4th Int. Conf. on Magnet Technology, AEC C0NF-720903:539; Sep 72.

Combination of a Power Transmission Line and an Active Track for a Magnetically Suspended, High Speed Train;

Clark, A. F.

J. Appl. Phys. 43(8):3598; Aug 72.

A Preliminary Investigation of the Behavior of High Purity Copper in High

Magnetic Fields;

Fickett, F. R. INCRA Annual Report, INCRA Project No. 186; Jun 72.

Standard Reference Materials: The Eddy Current Decay Method for Resistivity

Characterization of High Purity Metals;

Clark, A. F.; Deason, V. A.; Hust, J. G. ; Powell, R. L.

NBS SP 260-39; May 72.

Defect Annealing (4 to 295 K) After Martensitic Phase Transformation in an Fe

29 Ni Alloy;

Reed, R. P.; Clark, A. F.; Schramm, R. E. Scripta Met. 5:485; 71.

Longitudinal Magnetoresi stance Anomalies;

Fickett, F. R.; Clark, A. F.

J. Appl. Pys. 42:217; 71.

791
Magnetoresi stance of Very Pure Polycrystal 1 i ne Aluminum;

Fickett, F. R. Phys. Rev. B 3:1941; 71.

Martensitic Transformation Detection in Cryogenic Steels (Magnetometer

Devel opment)

Fickett, F. R. NBS TN 613; 71.

Characterization of High Purity Metals by the Eddy Current Decay Method;

Clark, A. F.; Deason, V. A.; Powell, R. L. Mater. Res. Stand. 11(8) :25-28; Aug 71.

Lorenz Ratio as a Tool for Predicting Thermal Conductivity of Metals and

Al loys;

Hust, J. G. ; Clark, A. F.

Mater. Res. Stand. 11 (8) :22-24; Aug 71.

Low Temperature Electrical Resistivity of Some Engineering Alloys;

Clark, A. F.; Childs, G. E.; Wallace, G. H. Adv. Cryo. Eng., New York: Plenum Press, 15:85-90; 70.

Low Temperature Specific Heat and Thermal Expansion of Alloys;

Clark, A. F.; Kropschot, R. H.

Int. Inst, of Refrigeration, Commission I, Sep 70, Tokyo, Bulletin de

l'Institut Internationale du Froid, Annexe 1970-2, p 249; 70.

Resistivity of Polycrystal line Aluminum and Copper in High Magnetic Fields

The Effect of Temperature and Purity;

Fickett F. R.

Appl. Phys. Lett. 17(12) :525-527; 70.

Low Temperature Electrical Resistivity of Some Engineering Alloys;

Clark, A. F.; Childs, G. E.; Wallace, G. H. Cryogenics 10:295-305; Aug 70.

792
APPENDIX III

793
i
A BIBLIOGRAPHY OF THE NBS ELECTROMAGNETIC FIELDS DIVISION PUBLICATIONS

Edited by

Kathryn A. Gibson and Charles K. S. Miller

This bibliography lists the publications of the personnel of the NBS

Electromagnetic Fields Division in the period from January 1982 through

December 1983.

Key words: antennas; electromagnetic interference; electromagnetic

metrology; microwaves; noise; non-ionizing radiation; radiation hazards;

waveform metrology.

INTRODUCTION

This bibliography updates and supplements NBSIR 80-1635 which lists the

publications of the Electromagnetic Fields Division from January 1970 through

December 1979, and NBSIR 82-1673 which lists publications from January 1980

through December 1981. The publications listed in this supplement were published in 1982 and 1983.

The Electromagnetic Fields Division was formed during the reorganization

of NBS in April 1978. The Division develops measurement methods and standards, and provides metrological support for: antenna systems, noise measurement equipment, electromagnetic interference, electromagnetic environmental

characterization equipment especially for near fields to sources or a near-field like condition, electromagnetic emission and immunity (or susceptibility) testing equipment, and waveform metrology. The editors have attempted to include all work published by the staff members of the Division, in

the period from January 1982 through December 1983. There are a few exceptions. Relevant work performed by authors prior to joining this Division or work by authors who have left this Division has been included because of its

special significance. A miscellaneous category includes general publications.

A few references are listed in more than one category where appropriate.

There are several other information sources that may be useful to the

reader who is interested in activities at NBS connected with electromagnetic

metrology. Companion bibliographies list publications of the Electromagnetic

Technology Division. Its topics include metrology for: microwave and

millimeter wave circuits, laser systems, optical communication equipment,

systems using transient or pulsed electromagnetic phenomena, cryoelectronics,

superconductors and other unusual electrical engineering materials. Three

bibliographies of the publications of the former Electromagnetics Division

have been published: NBSIR 73-320 (July 1972-June 1973); NBSIR 74-395 (July

1973-June 1974); NBSIR 75-818 (July 1974-June 1975). These were preceded by a series of unpublished reports (edited by H. M. Altschuler) that cover the

period back to 1956. An excellent summary of the whole field of

electromagnetic metrology as it stood in 1967 was published as a special issue

of the IEEE Proceedings (volume 55, June 1967). Advances in the following

decade were described in another special issue of the same journal (volume 66,

April 1978).

795
A NOTE ON ABBREVIATIONS

Most readers will be familiar with the commonly used abbreviations for

the names of the various professional journals that appear in this bibliography. There are also several publication series that are peculiar to NBS and

call for explanation. They are:

NBSIR - NBS Interagency/Internal Report

NBS-TN - NBS Technical Note

NBS-SP - NBS Special Publication

NBS-MN - NBS Monograph

Purchase Procedures and Document Availability

NBS Technical Notes and Special Publications may be purchased from the

Superintendent of Documents, U.S. Government Printing Office, Washington, DC

20402. Orders must be accompanied by postal money order, express money order,

or check made out to the Superintendent of Documents.

NBS Interagency/Internal Reports (NBSIR* s) may be purchased from the

National Technical Information Service, Springfield, VA 22161. Order must be

accompanied by postal money order, express money order, or check made out to

the NTIS.

Reprints of papers published in non-NBS media may be available in limited

quantities from the authors.

796
Antennas

The Orbiting Standards Package: A Recal ibratable Satellite Instrument Assembly

for Measuring Large Earth Station Antennas

Baird, R. C. and Estin, A. J.

Proc, Antenna Measurement Techniques Assoc., New Mexico State

University, Las Cruces, NM, Oct. 5-7, 1982, pp. 5-1 - 5-9

October 1982

Handbook for Broadband Isotropic Antenna System, Volume 1 - Operations Manual

Bensema, W. D. NBSIR 83-1693

July 1983

Optimized Wavelength-Sized Scalar Horns as Antenna Radiation Standards

Estin, A. J.; Stubenrauch, C. F.; Repjar, A. G.; and Newell, A. C.

IEEE Trans, on Instrumentation and Measurement

Vol. IM-31, No. 1, pp. 53-56

March 1982

A Partial Loop Source of E & H Fields for Antenna Factor Calibration (A Loop

Cell)

FitzGerrell, R. G.

Proc, Antenna Measurement Techniques Assoc., New Mexico State

University, Las Cruces, NM, Oct. 5-7, 1982, pp. 15-1 - 15-22

October 1982

Computations of Antenna Side-Lobe Coupling in the Near Field Using Approximate

Far-Field Data

Francis, M. H. and Yaghjian, A. D.

NBSIR 82-1674

August 1982

The Characteristics of Iris-Fed Millimeter Wave Rectangular Microstrip Patch

Antennas

Greenlee, D. H.; Kanda, M.; and Chang, D. C.

NBS Tech Note 1063

October 1983

The Characteristics of Iris-Fed Millimeter Wave Rectangular Microstrip Patch

Antennas

Kanda, M.; Chang, D. C; and Greenlee, D. H.

Proc, 1982 IEEE AP/S Symp., University of New Mexico, Albuquerque,

NM, May 24-28, 1982, pp. 292-295

May 1982

The Effects of Resistive Loading on TEM Horns

Kanda, M.

IEEE Trans, on EMC

Vol. EMC-24, No. 2, pp. 245-255

May 1982

797
Antennas (Cont'd)

Antenna Gain Measurements by an Extended Version of the NBS Extrapolation

Method

Repjar, A. G.; Newell, A. C; and Baird, R. C.

IEEE Trans, on Instrumentation and Measurement

Vol. IM-32, No. 1, pp. 88-91

March 1983

Antenna Gain Measurements by an Extended Version of the NBS Extrapolation

Method

Repjar, A. G.; Newell, A. C; and Baird, R. C.

Digest, Conference on Precision Electromagnetic Measurements,

Boulder, CO, June 28-July 1, 1982, pp. F-7 - F-9

June 1982

Accurate Evaluation of a Millimeter Wave Compact Range Using Planar Near-Field

Scanning

Repjar, A. G. and Kremer, D. P. IEEE Trans, on Antennas and Propagation

Vol. AP-30, No. 3, pp. 419-425

May 1982

International Intercomparison of Electric Field Strength at 100 MHz

Stubenrauch, C. F.; Spiess, W.; Galliano, P. G.; and Babij, T.

IEEE Trans, on Instrumentation and Measurement

Vol. IM-32, No. 1, pp. 235-237

March 1983

International Intercomparison of Electric Field Strength at 100 MHz

Stubenrauch, C. F.; Spiess, W.; Galliano, P. G.; and Babij, T.

Digest, Conference on Precision Electromagnetic Measurements,

Boulder, CO, June 28-July 1, 1982, pp. P-3 - P-4

June 1982

Precision Measurement of Antenna System Noise Using Radio Stars

Wait, D. F. IEEE Trans, on Instrumentation and Measurement

Vol. IM-32, No. 1, pp. 110-116

March 1983

Precision Measurement of Antenna System Noise Using Radio Stars

Wait, D. F. Digest, Conference on Precision Electromagnetic Measurements,

Boulder, CO, June 28-July 1, 1982, p. F-17

June 1982

Input Impedance of a Probe Antenna Exciting a TEM Cell

Wilson, P. F.; Chang, D. C; and Ma, M. T.

NBS Tech Note 1054

April 1982

798
Antennas (Cont'd)

Optical Modulator and Link for Broadband Antennas

Wyss, J. C; Kanda, M. ; Melquist, D. G.; and Ondrejka, A. R.

Digest, Conference on Precision Electromagnetic Measurements,

Boulder, CO, June 28-July 1, 1982, pp. P-16 - P-17

June 1982

Approximate Formulas for the Far Field and Gain of Open-Ended Rectangular

Waveguide

Yaghjian, A. D.

NBSIR 83-1689

May 1983

A Delta-Distribution Derivation of the Electric Field in the Source Region

Yaghjian, A. D.

Journal of the Electromaqnetics Society

Vol. 2, No. 2, pp. 161-167

April -June 1982

Efficient Computation of Antenna Coupling and Fields Within the Near-Field

Region

Yaghjian, A. D.

IEEE Trans, on Antennas and Propagation

Vol. AP-30, No, 1, pp. 113-128

January 1982

799
Electromagnetic Interference

Measurement of Electromagnetic Radiation from Electric Rail Cars

Adams, J. W.

NBSIR 82-1669

August 1982

Digest, 1982 Conference on Precision Electromagnetic Measurements, Boulder,

CO, June 28-July 1, 1982

Alspach, W. J. (Editor)

June 1982

Planning Guidance for Future EMI Measurement Instrumentation

Arthur, M. G.; Orr, R. D.; and Reeve, G. R.

NBSIR 82-1662

April 1982

Handbook for Broadband Isotropic Antenna System Volume 1 - Operations Manual

Bensema, W. D.

NBSIR 83-1693

July 1983

Comparison of Open-Field, Anechoic Chamber and TEM Cell Facilities/Techniques

for Performing Electromagnetic Radiated Emissions Measurements

Crawford, M. L.

Record, IEEE 1983 International Symposium on Electromagnetic Compatibility, Arlington, VA, August 23-25, 1983, pp. 413-418

August 1983

Evaluation of Shielded Enclosure for EMI/EMC Measurements Without and With RF

Anechoic Material

Crawford, M. L.

Proceedings, 1983 EMC Symposium and Exhibition, Zurich, Switzerland,

March 8-10, 1983, pp. 397-402

March 1983

Improving the Repeatability of EM Susceptibility Measurements of Electronic

Components When Using TEM Cells

Crawford, M. L.

SAE Technical Paper Series, 830607, International Congress and

Exposition, Detroit, MI, February 28-March 4, 1983, pp. 1-8

March 1983

Development of the NBS Isotropic Magnetic-Field Meter (MFM-10), 300 kHz to 100

MHz

Driver, L. D. and Cruz, J. E. Proceedings, 1982 International Symposium on EMC, Santa Clara, CA,

September 8-10, 1982, pp. 460-467

September 1982

800
Electromagnetic Interference (Cont'd)

E-Fields Over Ground

FitzGerrell , R. G.

Record, IEEE 1983 International Symposium on Electromagnetic Compatibility, Arlington, VA, August 23-25, 1983, pp. 6-9

August 1983

A Partial Loop Source of E & H Fields for Antenna Factor Calibration (A Loop

Cell)

FitzGerrell , R. G.

Proceedings, Antenna Measurement Techniques Association, New Mexico

State University, Las Cruces, NM, October 5-7, 1982, pp. 15-1 - 15-22

October 1982

Free Space Transmission Loss for Anechoic Chamber Performance Evaluation

FitzGerrell, R. G.

IEEE Transactions on Electromagnetic Compatibility, Vol. EMC-24, No.

3, pp. 356-358

August 1982

Time Domain Sensors for Radiated Impulsive Measurements

Kanda, M. and Ries, F. X.

IEEE Transactions on Antennas and Propagation, Vol. AP-31, No. 3,

pp. 438-444

May 1983

An Electric and Magnetic Field Sensor for Simultaneous Electromagnetic

Near-Field Measurements - Theory

Kanda, M.

NBS Tech Note 1062

April 1983

An Electric and Magnetic Field Sensor Concept for Simultaneous Near-Field

Electromagnetic Components When Using TEM Cells

Kanda, M.; Ries, F. X.; Driver, L. D.; and Orr, R. D.

Proceedings 1983 Electromagnetic Compatibility Symposium and Exhibition, Zurich, Switzerland, March 8-10, 1983, pp. 263-266

March 1983

Time Domain Sensors for Radiated Impulsive Measurements

Kanda, M. and Ries, F. X.

Proceedings, IEEE 1982 International Symposium on Electromagnetic

Compatibility, Santa Clara, CA, September 8-10, 1982, pp. 296-301

September 1982

801
Electromagnetic Interference (Cont'd)

Design Considerations for Broadband Magnetic-Field Sensors

Kanda, M.; Ries, F. X.; Driver, L. D.; and Orr, R. D.

Digest, Conference on Precision Electromagnetic Measurements,

Boulder, CO, June 28-July 1, 1982, pp. P-ll - P-13

June 1982

The Effects of Resistive Loading on TEM Horns

Kanda, M.

IEEE Transactions on EMC

Vol. EMC-24, No. 2, pp. 245-255

May 1982

A New Method for Determining the Emission Characteristics of an Unknown

Interference Source

Koepke, G. H. and Ma, M. T.

Proceedings, 1983 Electromagnetic Compatibility Symposium and

Exhibition, Zurich, Switzerland, March 8-10, 1983, pp. 263-266

March 1983

A New Method for Determining the Emission Characteristics of an Unknown

Interference Source

Koepke, G. H. and Ma, M. T.

Proceedings, IEEE 1982 International Symposium on Electromagnetic

Compatibility, Santa Clara, CA, September 8-10, 1982, pp. 151-156

September 1982

Design Consideration of Reverberating Chambers for Electromagnetic Interference Measurements

Liu, B.-H.; Chang, D. C; and Ma, M. T.

Record, IEEE 1983 International Symposium on Electromagnetic Compatibility, Arlington, VA, August 23-25, 1983, pp. 508-512

August 1983

Eigenmodes and the Composite Quality Factor of a Reverberating Chamber

Liu, B.-H.; Chang, D. C; and Ma, M. T.

NBS Tech Note 1066

August 1983

Uncertainties in Extracting Radiation Parameters for an Unknown Interference

Source Based on Power and Phase Measurements

Ma, M. T. and Koepke, G. H.

NBS Tech Note 1064

June 1983

802
Electromagnetic Interference (Cont'd)

A Method to Quantify Radiation Characteristics of an Unknown Interference

Source

Ma, M. T. and Koepke, 6. H. NBS Tech Note 1059

October 1982

A Study of Distribution of Electromagnetic Fields Inside Buildings with

Apertures Excited by an External Source

Ma, M. T. and Arthur, M. G.

NBSIR 82-1659

February 1982

The EMI Measurement Challenge

Miller, C. K. S.

Proceedings, Measurement Science Conference, Palo Alto, CA, January

20-21, 1983, pp. 189-197

May 1983

International Intercomparison of Electric Field Strength at 100 MHz

Stubenrauch, C. F.; Spiess, W.; Galliano, P. G.; and Babij, T.

IEEE Trans, on Instrumentation and Measurement

Vol. IM-32, No. 1, pp. 235-237

March 1983

Radiated EMI Instrumentation Errors

Taggart, H. E. EMC Technology Magazine

Volume 1, No. 4, pp. 26-35

October 1982

Theoretical and Experimental Analysis of Coupling Characteristics of Dual TEM

Cells

Wilson, P. F.; Chang, D. C; Ma, M. T.; and Crawford, M. L.

Record, IEEE 1983 International Symposium on Electromagnetic Compatibility, Arlington, VA, August 23-25, 1983, pp. 513-517

August 1983

Input Impedance of a Probe Antenna Exciting a TEM Cell

Wilson, P. F.; Chang, D. C; and Ma, M. T.

NBS Tech Note 1054

April 1982

Optical Modulator and Link for Broadband Antennas

Wyss, J. C; Kanda, M.; Melquist, D. G.; and Ondrejka, A. R.

Digest, Conference on Precision Electromagnetic Measurements,

Boulder, CO, June 28-July 1, 1982, pp. P-16 - P-17

June 1982

803
Noise

Design and Error Analysis for the WR10 Thermal Noise Standard

Daywitt, W. C.

NBS Tech Note 1071

December 1983

Preliminary Examination of 20 GHz G/T Measurements of Earth Terminals

Wait, D. F. and Daywitt, W. C.

NBSIR 83-1686

March 1983

Precision Measurement of Antenna System Noise Using Radio Stars

Wait, D. F. IEEE Transactions on Instrumentation and Measurement

Vol. IM-32, No. 1, pp. 110-116

March 1983

Earth Terminal Measurement System Operations Manual (Revised)

Wait, D. F. NBSIR 83-1679

January 1983

Precision Measurement of Antenna System Noise Using Radio Stars

Wait, D. F. Digest, Conference on Precision Electromagnetic Measurements,

Boulder, CO, June 28-July 1, 1982, pp. F-17

June 1982

804
Waveform Metrology

Reference Flat Pulse Generator

Andrews, J. R.; Bell, B. A.; and Baldwin, E. E.

NBS Tech Note 1067

October 1983

Reference Waveform Flat Pulse Generator

Andrews, J. R.; Bell, B. A.; Nahman, N. S.; and Baldwin, E. E. IEEE Transactions on Instrumentation and Measurement

Vol. IM-32, No. 1, pp. 27-32

March 1983

Status of Reference Waveform Standards Development at NBS

Andrews, J. R.; Nahman, N. S.; and Bell, B. A.

NBS Special Publication 634, Proceedings of the Seminar on Waveform

Recorder Measurement Needs and Techniques for Evaluation/Calibration, Boulder, CO, October 15, 1981, pp. 69-88

June 1982

The Measurement and Deconvolution of Time Jitter in Equivalent-Time Waveform

Samplers

Gans, W. L.

IEEE Transactions on Instrumentation and Measurement

Vol. IM-32, No. 1, pp. 126-133

March 1983

Photoconductive Switches Used for Waveform Generation at the National Bureau

of Standards

Lawton, R. A.

Proceedings, Picosecond Optoelectronics Conference, SPIE - The

International Society for Optical Engineering, San Diego, CA, August

24-26, 1983

Vol. 439, pp. 88-94

August 1983

Performance Standards for Waveform Recorders

Lawton, R. A.

IEEE Transactions on Nuclear Science

Vol. NS-30, No. 1, pp. 263-266

February 1983

Precision Picosecond-Microsecond Electromagnetic Waveform Measurement at NBS

Lawton, R. A.

NBS Special Publication 628

pp. 392-407

June 1982

805
Waveform Metrology (Cont'd)

Proceedings of the Seminar on Waveform Recorder Measurement Needs and Techniques for Evaluation/Calibration

Lawton, R. A. (Editor)

NBS Special Publication 634, Proceedings of the Seminar on Waveform

Recorder Measurement Needs and Techniques for Evaluation/Calibration, Boulder, CO, October 15, 1981,

June 1982

Picosecond Domain Waveform Measurements; Status and Future Directions

Nahman, N. S.

IEEE Transactions on Instrumentation and Measurement

Vol. IM-32, No. 1, pp. 117-124

March 1983

Some Generic Waveform Recorder Problems

Nahman, N. S.

NBS Special Publication 634, Proceedings of the Seminar on Waveform

Recorder Measurement Needs and Techniques for Evaluation/Calibration, Boulder, CO, October 15, 1981, pp. 1-5

June 1982

806
Mi seel laneous

Electromagnetic Modeling of Oil Shale Retorts for Remote Sensing Purposes

Chew, H.

IEEE Transactions on Geoscience and Remote Sensing

Vol. GE-20, No. 4, pp. 510-517

October 1982

Bibliography of the Electromagnetic Fields Division Publications

Gibson, K. A. and Miller, C. K. S.

NBSIR 82-1673

August 1982

Book Review - Leaky Feeders and Subsurface Radio Communications

Hill, D. A

-^

IEEE Antennas and Propagation Society Newsletter

Vol. 25, No. 3, pp. 23-24

June 1983

Dielectric Measurements of Oil Shale as Functions of Temperature and Frequency

Jesch, R. L.

NBSIR 83-1683

January 1983

Improved Coal Interface Detector

Roe, K. C. and Wittmann, R. C.

NBSIR 82-1663

May 1982

Acoustical Interferometer Wader AIW Final Report of Advanced Development Test

and Evaluation

Stoltenberg, R. E. and Wittmann, R. C.

NBSIR 82-1671 (restricted)

June 1982

807
A BIBLIOGRAPHY OF THE NBS ELECTROMAGNETIC FIELDS DIVISION PUBLICATIONS

Edited by

Kathryn A. Gibson and Charles K. S. Miller

This bibliography lists the publications of the personnel of the NBS

Electromagnetic Fields Division in the period from January 1980 through

December 1981.

Key words: antennas; electromagnetic interference; em material

properties; electromagnetic metrology; microwaves; noise; non-ionizing

radiation; radiation hazards.

INTRODUCTION

This bibliography updates and supplements NBSIR 80-1635 which lists the

publications of the Electromagnetic Fields Division from January 1970 to

December 1979. The publications listed in this supplement were published in

1980 and 1981.

The Electromagnetic Fields Division was formed during the reorganization

of NBS in April 1978. The Division develops measurement methods and

standards, and provides metrological support for: antenna systems, noise

measurement equipment, electromagnetic interference, electromagnetic

environmental characterization equipment especially for near fields to sources

or a near-field like condition, electromagnetic emission and immunity (or

susceptibility) testing equipment, and for measuring equipment used for

dielectric or loss characterization of materials. The editors have attempted

to include all work published by the staff members of the Division, in the

period from January 1980 through December 1981. There are a few exceptions.

Relevant work performed by authors prior to joining this Division or work by

authors who have left this Division has been included because of its special

significance. A miscellaneous category includes general publications. A few

references are listed in more than one category where appropriate.

There are several other information sources that may be useful to the

reader who is interested in activities at NBS connected with electromagnetic

metrology. Companion bibliographies list publications of the Electromagnetic

Technology Division. Its topics include metrology for: microwave circuits,

laser systems, optical communication equipment, systems using transient or pulsed electromagnetic phenomena, cryoelectronics, superconductors and other

unusual electrical engineering materials. Three bibliographies of the

publications of the former Electromagnetics Division have been published:

NBSIR 73-320 (July 1972-June 1973); NBSIR 74-395 (July 1973-June 1974); NBSIR

75-813 (July 1974-June 1975). These were preceded by a series of unpublished

reports (edited by H. M. Altschuler) that cover the period back to 1956. An

excellent summary of the whole field of electromagnetic metrology as it stood

in 1967 was published as a special issue of the IEEE Proceedings (volume 55,

June 1967). Advances in the following decade were described in another

special issue of the same journal (volume 66, April 1978).

808
A NOTE ON ABBREVIATIONS

Most readers will be familiar with the commonly used abbreviations for

the names of the various professional journals that appear in this

bibliography. There are also several publication series that are peculiar to

NBS and call for explanation. They are:

NBSIR - NBS Interagency/Internal Report

NBS-TN - NBS Technical Note

NBS-SP - NBS Special Publication

NBS-MN - NBS Monograph

Purchase Procedures and Document Availability

NBS Technical Notes and Special Publications may be purchased from the

Superintendent of Documents, U.S. Government Printing Office, Washington,

D.C. 20402. Orders must be accompanied by postal money order, express money

order, or check made out to the Superintendent of Documents.

NBS Interagency/Internal Reports (NBSIR' s) may be purchased from the

National Technical Information Service, Springfield VA 22161. Order must be

accompanied by postal money order, express money order, or check made out to

the NTIS.

Reprints of papers published in non-NBS media may be available in limited

quantities from the authors.

809
Antennas

Microwave Antenna Measurement Services at the National Bureau of Standards

Baird, R. C. Digest, Antenna Meas. Tech. Assn. Mtg.

Oct. 1981

Broadband Orthogonal Array Antenna System: Microprocessor Control and

Computation

Bensema, W. D.

Record, EMC '81

Aug. 1981

Optimized Wavelength-Sized Scalar Horns as Antenna Radiation Standards

Estin, A. J.; Stubenrauch, C. F.; Repjar, A. G.; and Newell, A. C.

1981 EEMTIC Digest

June 1981

The Characteristics of a Linear Antenna with Tapered Resistive and Capacitive

Loading

Kanda, M.

Proc, IEEE AP Symp., Quebec, Canada

June 2-6, 1980, Vol. 2, AP.18-2, pp 696-699

The Time Domain Characteristics of a Traveling Wave Linear Antenna with Linear

and Non-Linear Loads

Kanda, M.

IEEE Trans. Ant. & Prop.

March 1980, Vol. AP-28, No. 2

Analytical and Numerical Techniques for Analyzing an Electrically Short Dipole

with a Nonlinear Load

Kanda, M.

IEEE Trans. Ant. & Prop.

Jan. 1980, Vol. AP-28, No. 1, pp 71-78

Transients in Resistively Loaded Antennas and Their Comparison with Conical

Antennas and a TEM Horn

Kanda, M.

IEEE Trans. Ant. & Prop.

Jan. 1980, Vol. AP-28, No. 1, pp 132-136

Plane-Wave Scattering-Matrix Theory of Antennas and Antenna-Antenna

Interactions

Kerns, D. M.

NBS Monograph 162

June 1981

Spherical -Wave Source-Scattering-Matrix Analysis of the Mutual Coupling

Between Two Antennas

Lewis, R. L. Digest, 1981 IEEE/AP-S Int'l Symp.

June 1981

810
Antennas (Cont'd)

Efficient Computation of the Far Field Radiated by an Arbitrary RectangularAperture Distribution

Lewis, R. L.

NBSIR 81-1643

April 1981

Synthesis of Broadband Antenna Arrays as Possible Over-the-Horizon Radars

Ma, M. T.

Book, "Research Topics in EM Wave Theory" (John Wiley Interscience)

1981, Chapter 9

Synthesized Isotropic Pattern Antennas for EM Field Measurements

Reeve, G. R. Symp. Record, "EMC '81," 1981 IEEE Int'l Symp. on Electromagnetic

Compatibility, Boulder, CO, Aug. 18-20, 1981

Aug. 1981

A Frequency Tracking, Tuned, Receiving Monopole

Reeve, G. R. and Wainwright, A. E.

Record, IEEE/APS Symp.

June 1981

Results of Planar Near-Field Measurements on a Compact Range at 18 and 54 GHz

Repjar, A. G. and Kremer, D. P.

Digest, 1980 IEEE/AP-S Symposium

Determination of Mutual Coupling Between Co-Sited Microwave Antennas and

Calculation of Near-Zone Electric Field

Stubenrauch, C. F. and Yaghjian, A. D.

NBSIR 81-1630

June 1981

Fields of a Horizontal Loop of Arbitrary Shape Buried in a Two-Layer Earth

Wait, J. R. and Hill, D. A.

Radio Science

Sep. -Oct. 1980, Vol. 15, No. 5, pp 903-912

A Delta-Distribution Derivation of the Electric Field in the Source Region

Yaghjian, A. D.

Digest, 1981 IEEE/AP-S Int'l Symp.

June 1981

An Approximate Expression. for the Principal Beamwidthsof Directive Antennas in

Terms of Aperture Fields

Yaghjian, A. D.

NBSIR 81-1644

March 1981

811
Electromagnetic Interference

Broadband Orthogonal Array Antenna System: Microprocessor Control and

Computation

Bensema, W. D.

Record, EMC '81

Aug. 1981

A Temperature Probe for Radio-Frequency Heated Material

Bowman, R. R.

NBSIR 81-1634

Jan. 1981

Requirements for an Effective National Nonionizing Radiation Measurement

System

Clark, H. E.

NBS Special Pub. 613

June 1981

Options to Open-Field and Shielded Enclosure Electromagnetic Compatibility

Measurements

Crawford, M. L.

Proc, Int'l EMC Symp., Zurich, Switzerland, March 10-12, 1981

March 1981

Evaluation of a Reverberation Chamber Facility for Performing EM Radiated

Fields Susceptibility Measurements

Crawford, M. L.

NBSIR 81-1638

Feb. 1981

Predicting Free-Space Radiated Emissions from Electronic Equipment Using TEM

Cell and Open-Field Site Measurements

Crawford, M. L. and Workman, J. L.

Record, 1980 Int'l EMC Symposium

Spherical Dipole for Radiating Standard Fields

Crawford, M. L. and Workman, J. L.

Digest 1980 CPEM Conf.

June 1980

Design, Construction, and Calibration of the Broadband Electric Field Monitor

(EFM-5)

Cruz, J. E.

Symp. Record, "EMC '81," 1981 IEEE Int'l Symp. on Electromagnetic

Compatibility, Boulder, CO, Aug. 18-20, 1981

Aug. 1981

Free-Space Transmission Loss for Anechoic Chamber Performance Evaluation

FitzGerrell, R. G.

Symp. Record, "EMC '81," 1981 IEEE Int'l Symp. on Electromagnetic

Compatibility, Boulder, CO, Aug. 18-20, 1981

Aug. 1981

812
Electromagnetic Interference (Cont'd)

Electromagnetic Characteristics of a Coaxial Cable with Periodic Slots

Hill , D. A. and Wait, J. R. IEEE Trans, on EMC

Nov. 1980, Vol. EMC-22, No. 4, pp 303-307

Propagation Along a Coaxial Cable with a Helical Shield

Hill , D. A. and Wait, J. R. IEEE Trans, on MTT

Feb. 1980, Vol. MTT-28, No. 2, pp 84-89

A Broadband, Isotropic, Real-Time, Electric-Field Sensor (BIRES) Using

Resistively Loaded Di poles

Kanda, M. ; Ries, F. X.; and Belsher, D. R.

IEEE Trans. EMC

Aug. 1981, Vol. EMC-23, No. 3

Theoretical and Experimental Investigations of Electromagnetic Field

Distortion Due to a Perfectly Conducting Rectangular Cylinder in a Transverse

Electromagnetic Cell

Kanda, M.

NBS Tech. Note 1028

April 1981

Theoretical and Experimental Investigations of Loading Effects Due to a Perfectly Conducting Rectangular Cylinder in a Transverse Electromagnetic Cell

Kanda, M.

4th Symp. and Tech. Exhib. on EMC, Zurich, Switzerland

March 1981

The Characteristics of a Linear Antenna with Tapered Resistive and Capacitive

Loading

Kanda, M.

Proc, IEEE AP Symp., Quebec, Canada

June 2-6, 1980, Vol. 2, AP.18-2, pp 696-699

The Time Domain Characteristics of a Traveling Wave Linear Antenna with Linear

and Non-Linear Loads

Kanda, M.

IEEE Trans. Ant. and Prop.

March 1980, Vol. AP-28, No. 2

Analytical and Numerical Techniques for Analyzing an Electrically Short Dipole

with a Nonlinear Load

Kanda, M.

IEEE Trans. Ant. and Prop.

Jan. 1980, Vol. AP-28, No. 1, pp 71-78

Transients in Resistively Loaded Antennas and Their Comparison with Conical

Antennas and a TEM Horn

Kanda, M.

IEEE Trans. Ant. and Prop.

Jan. 1980, Vol. AP-28, No. 1, pp 132-136

813
Electromagnetic Interference (Cont'd)

Background and Present Status of RF Probe Development at NBS

Larsen, E. B. Symp. Record, "EMC '81," 1981 IEEE Int'l Symp. on Electromagnetic

Compatibility, Boulder, CO, Aug. 18-20, 1981

Aug. 1981

Design and Calibration of the NBS Isotropic Electric-Field Monitor (EFM-5),

0.2 to 1000 MHz

Larsen, E. B. and Ries, F. X. NBS Tech. Note 1033

March 1981

A Method of Determining the Emission and Susceptibility Levels of Electrically

Small Objects Using a TEM Cell

Ma, M. T.; Chang, D. C. ; and Sreenivasiah, I. NBS Tech. Note 1040

April 1981

Excitation of a TEM Cell by a Vertical Electric Hertzian Dipole

Ma, M. T.; Wilson, P. F.; and Chang, D. C. NBS Tech. Note 1037

March 1981

Challenges of EMI Measurements

Miller, C. K. S.

Proc. NCSL Conf., Sep. 22-25, 1980

Oct. 1980

Synthesized Isotropic Pattern Antennas for EM Field Measurements

Reeve. G. R.

Symp. Record, "EMC '81," 1981 IEEE Int'l Symp. on Electromagnetic

Compatibility, Boulder, CO, Aug. 18-20, 1981

Aug. 1981

A Frequency Tracking, Tuned, Receiving Monopole

Reeve, G. R. and Wainwright, A. E.

Record, IEEE/APS Symp.

June 1981

Influence of Electromagnetic Interference on Electronic Devices

Ries, F. X. and Miller, C. K. S.

Bulletin, OIML Seminar

Dec. 1981

A Critical Study of Emission and Susceptibility Levels of Electrically Small

Objects from Tests Inside a TEM Cell

Sreenivasiah, I.; Chang, D. C. ; and Ma, M. T.

Record, EMC '81 Symp.

Aug. 1981

814
Electromagnetic Interference (Cont'd)

Emission Characteristics of Electrically Small Radiating Sources from Tests

Inside a TEM Cell

Sreeni vasiah, I.; Chang, D. C; and Ma, M. T.

IEEE Trans. EMC

Aug. 1981, Vol. EMC-23, No. 3

Characterization of Electrically Small Radiating Sources by Tests Inside a Transmission Line Cell

Sreeni vasiah, I.; Chang, D. C; and Ma, M. T.

NBS Tech. Mote 1017

March 1980

Methods of Suppressing Automotive Interference

Taggart, H. E.

NBS Special Pub. 480-44

Nov. 1981

815
Noise

NBS 30/60 Megahertz Noise Measurement System Operation and Service Manual

Counas, G. J. and Bremer, T. H.

NBSIR 81-1656

Dec. 1981

Evaluation of Signal Plus Noise Detection Error in an Envelope Detector with

Logarithmic Compression

Estin, A. J. and Daywitt, W. C.

IEEE Trans, Info. Theory

Sep. 1981, Vol. IT-27, No. 5, pp 663-664

Evaluation of Signal Plus Noise Detection Error in an Envelope Detector with

Logarithmic Compression

Estin, A. J. and Daywitt, W. C.

NBSIR 79-1614 (R)

Aug. 1979

Addendum to Earth Terminal Measurement System Maintenance Manual

Wakefield, J. P.

NBSIR 81-1641

Oct. 1981

816
Properties of Materials

Dielectric Measurements in a Shielded Open Circuit Coaxial Line

Bussey, H. E.

IEEE Trans. Inst, and Meas.

June 1980, Vol. IM-29, No. 2

Active Microwave Water Equivalence Measurements

Ellerbruch, D. A. and Boyne, H. S.

Proc, Ft. Collins Snow Workshop

1980

817
Mi seel 1 aneou

A Temperature Probe for Radio-Frequency Heated Material

Bowman, R. R. NBSIR 81-1634

Jan. 1981

Modeling of Oil Shale Retorts for Electromagnetic Sensing Techniques

Chew, H.

NBSIR 81-1653

Nov. 1981

Requirements for an Effective National Nonionizing Radiation Measurement

System

Clark, H. E.

NBS Special Pub. 613

June 1981

A Wideband RF Voltage Comparator

Driver, L. D. and Ries, F. X.

Digest, 1980 CPEM Conf.

June 1980

HF Ground Wave Propagation over Mixed Land, Sea, and Sea-Ice Paths

Hill , D. A. and Wait, J. R.

IEEE Trans, on Geoscience and Remote Sensing

Oct. 1981, Vol. GE-19, No. 4, pp 210-216

HF Radio Wave Transmission over Sea Ice and Remote Sensing Possibilities

Hill , D. A. and Wait, J. R.

IEEE Trans, on Geoscience and Remote Sensing

Oct. 1981, Vol. GE-19, No. 4, pp 204-209

HF Ground Wave Propagation over Sea Ice for a Spherical Earth Model

Hill , D. A. and Wait, J. R.

IEEE Trans, on Ant. and Prop.

May 1981, Vol. AP-19, No. 3, pp 525-527

Electromagnetic Characteristics of a Coaxial Cable with Periodic Slots

Hill , D. A. and Wait, J. R.

IEEE Trans, on EMC

Nov. 1980, Vol. EMC-22, No. 4, pp 303-307

On the Excitation of the Zenneck Surface Wave Over the Ground at 10 MHz

Hill , D. A. and Wait, J. R.

Annales des Telecommunications

May-June 1980, Vol. 35, Nos. 5-6, pp 1-4

Ground Wave Attenuation Function for a Spherical Earth with Arbitrary Surface

Impedance

Hill , D. A. and Wait, J. R. Radio Science

May-June 1980, Vol. 15, No. 3, pp 637-643

818
Miscellaneous (Cont'd)

Electromagnetic Theory of the Loosely Braided Coaxial Cable: Part II - Numerical Results

Hill , D. A. and Wait, J. R. IEEE Trans, on MTT

April 1980, Vol. MTT-28, No. 4, pp 326-331

Propagation Along a Coaxial Cable with a Helical Shield

Hill , D. A. and Wait, J. R. IEEE Trans, on MTT

Feb. 1980,, Vol. MTT-28, No. 2, pp 84-89

Evaluation of Three-Terminal and Four-Terminal Pair Capacitors at High

Frequencies

Jones, R. N. NBS Tech. Note 1024

Sep. 1980

Spherical Scanning Data Processing: An Algorithm for Halving the Data

Processing Effort when the Radiation into the Back Hemisphere is Negligible

Lewis, R. L. Digest, 1981 IEEE AP-S Int'l Symp.

June 1981

A Bibliography of Publications in the NBS Electromagnetic Fields Division

Miller, C. K. S.

NBSIR 80-1635

Nov. 1980

Fields of a Horizontal Loop of Arbitrary Shape Buried in a Two-Layer Earth

Wait, J. R. and Hill , D. A.

Radio Science

Sep. -Oct. 1980, Vol. 15, No. 5, pp 903-912

Addendum to Earth Terminal Measurement System Maintenance Manual

Wakefield, J. P. NBSIR 81-1641

Oct. 1981

Laser Fluorescence Diagnostics of Lithium Vapor for Inertial Confinement

Appl i cations

Wyss, J. C; Stwalley, W. C. ; Koch, M. E.; and Verma, K. K.

Topical Meeting on Inertial Confinement Fusion, San Diego, CA

Feb. 26, 1980

A Delta-Distribution Derivation of the Electric Field in the Source Region

Yaghjian, A. D.

Digest, 1981 IEEE/AP-S Symp.

June 1981

819
Miscellaneous (Cont'd)

Reply to Criticism on Electric Dyadic Functions in the Source Region

Yaghjian, A. D.

Proc, IEEE

Feb. 1981, Vol. 69, No. 2

Augmented Electric- and Magnetic-Field Integral Equations

Yaghjian, A. D.

Digest, Int'l URSI Symp. on EM Waves, Munich, Germany

Aug. 1980

Electric Dyadic Green's Functions in the Source Region

Yaghjian, A. D.

Proc, IEEE

Feb. 1980, Vol. 68, No. 2

820
A BIBLIOGRAPHY OF PUBLICATIONS IN THE NBS ELECTROMAGNETIC FIELDS DIVISION

Edited by

Charles K. S. Miller

This bibliography lists the publications of the personnel of the

NBS Electromagnetic Fields Division of NBS in the period from

January 1970 through December 1979. A few earlier references that

are directly related to the present work of the division are included.

Key words: Antennas; electromagnetic interference; em material

properties; electromagnetic metrology; microwaves; noise; non- ionizing radiation; radiation hazards;.

INTRODUCTION

The Electromagnetic Fields Division was formed during the reorganization

of NBS in April 1978. The Division develops measurement methods and

standards, and provides metrological support for: antenna systems, noise

measurement equipment, electromagnetic interference and radiation hazards

equipment, systems for measuring electromagnetic properties of materials, and

radiated electromagnetic fields and waves. For the individual staff members

of the Division, the reorganization brought a realignment of long term goals

but little immediate discontinuity in their work. It makes good sense therefore that this bibliography begin some time before the reorganization, so as to include the more recent origins of the present work of the Division.

The editor has attempted to include all work published by the staff members of

the Division, while they were employees of NBS, in the period from January

1970 through December 1979. There are a few exceptions. Work that is totally

unrelated to the present program has been excluded. Relevant work performed

by authors prior to joining this Division or work by authors now in other

parts of NBS has been included because of its special significance. A

miscellaneous category includes general publications. A few papers published

before 1970 have also been included because of their direct relationship to

the present program.

There are several other information sources that may be useful to the

reader who is interested in activities at NBS connected with electromagnetic

metrology. A companion bibliography: Metrology for Electromagntic

Technology, NBSIR 79-1625 (January 1980), lists the publications of the

Electromagnetic Technology Division. Its topics include metrology for:

antennas, satellite communications equipment, electromagnetic interference,

and hazards, and remote sensing. Three bibliographies of the publications of

the former Electromagnetic Division have been published: NBSIR-73-320 (July

1972-June 1973); NBSIR-74-395 (July 1973 - June 1974); NBSIR-75-818

(July 1974 - June 1975). These were preceded by a series of unpublished

reports (edited by H. M. Altschuler) that cover the period back to 1956. An

excellent summary of the whole field of electromagnetic metrology as it stood

in 1967 was published as a special issue of the IEEE Proceedings (volume 55,

June 1967). Advances in the following decade were described in another

special issue of the same journal (volume 66, April 1978).

821
A NOTE ON ABBREVIATIONS

Most readers will be familiar with the commonly used abbreviations for the names of the various professional journals that appear in this bibliography.

There are also several publication series that are peculiar to NBS and call

for explanation. They are:

Interagency/Internal Report

Technical Note

Special Publication

Monograph

Law Enforcement Standards Program Report

Office of Telecommunications Report

Purchase Procedures and Document Availability

NBS Technical Notes and Special Publications may be purchased from the

Superintendent of Documents, U.S. Government Printing Office, Washington, D.C.

20402. Orders must be accompanied by postal money order, express money order, or check made out to the Superintendent of Documents.

NBS Interagency/Internal Reports (NBSIR's) may be purchased from the

National Technical Information Service, Springfield VA 22161. Order must be

accompanied by postal money order, express money order, or check made out to

the NTIS.

Reprints of papers published in non-NBS media may be available in limited

quantities from the authors.

ACKNOWLEDGMENTS

A large part of the labor of preparing a bibliography is spent on collecting and arranging the material. I thank Frances Brown,

Kathryn A. Gibson, Susie Ann Rivera, and Susan Lee Rothschild for their

assistance with these chores. The prime source of material was the NTIS file,

with access through the Lockheed Dialog System. This was supplemented with

material supplied by the individual authors.

NBSIR - NBS

NBS-TN - NBS

NBS-SP - NBS

NBS-MN - NBS

LESP-RPT

OTR -

822
ANTENNA SYSTEMS METROLOGY

ANTENNA WITH INHERENT FILTERING ACTION;

BABIJ, TADEUSZ M. ; BOWMAN, RONALD R.; WACKER, PAUL F. PAT-APPL-590 355, PATENT-4 008 477, 6P 1975.

CALIBRATION OF BROADBEAM ANTENNAS USING PLANAR NEAR-FIELD

MEASUREMENTS;

CRAWFORD, M. L. CPEM DIGEST, P53-56 JUNE 28 - JULY 1, 1976

PLANAR NEAR-FIELD MEASUREMENTS ON PHASED ARRAY ANTENNAS;

NEWELL, ALLEN C; CRAWFORD, MYRON L. PROCEEDINGS OF INTERNATIONAL IEEE AP-S SYMPOSIUM, ATLANTA, GA. 10-12

JUNE 1974, PAPER 6-7 IN 1974 INTERNATIONAL IEEE/AP-S SYMPOSIUM

DIGEST, P423 JUNE 1974.

PLANAR NEAR-FIELD MEASUREMENTS ON HIGH PERFORMANCE ARRAY ANTENNAS;

NEWELL, ALLEN C; CRAWFORD, MYRON L.

NBSIR-74-380. lOlp JULY 1974.

TRANSMITTING AND RECEIVING FORMULAS FOR ANTENNAS AND FIELD INTENSITY METERS;

BOWMAN, RONALD R. 42p RADC-TR-68-445 TECHNICAL REPORT APRIL 1969.

A THEORETICAL STUDY OF UNBALANCED GROUND EFFECTS ON RECEIVING DIPOLES;

MA, M. T.

NBSIR-79-1605, 16p MAY 1979.

ANTENNAS AND THE ASSOCIATED TIME DOMAIN RANGE FOR THE MEASUREMENT OF IMPULSIVE

FIELDS;

LAWTON, ROBERT A.; ONDREJKA, ARTHUR R. NBS-TN-1008, 69p NOVEMBER 1978.

A RELATIVELY SHORT CYLINDRICAL BROADBAND ANTENNA WITH TAPERED

RESISTIVE LOADING FOR PICOSECOND PULSE MEASUREMENTS;

KANDA, MOTOHISA.

NBSIR-77-861. IEEE TRANS. ANTENNAS PROPAG. AP-26 N3 P439-447 MAY 1978.

TRANSIENTS IN RESISTIVELY LOADED ANTENNAS AND THEIR COMPARISON WITH CONICAL

ANTENNAS AND TEM HORNS;

KANDA, MOTOHISA.

NBSIR-78-876, 38p 1978.

G/T MEASUREMENT ERRORS WITH RADIO STARS;

DAYWITT, W. C; KANDA, M.

PROCEEDINGS OF IEEE INT. ANTENNAS AND PROPAGATION SYMP. URBANA/CHAMPAIGN, ILL.

2-4 JUNE 1975, SESSION 20, P460-463 1975.

ACCURACY CONSIDERATIONS IN THE MEASUREMENT OF THE POWER GAIN OF A

LARGE MICROWAVE ANTENNA;

KANDA, MOTOHISA.

PROCEEDINGS OF 1974 INTERNATIONAL IEEE/AP-S SYMPOSIUM,

ATLANTA, GA. 10-12 JUNE 1974, PAPER IN 1974 INTERNATIONAL IEEE/AP-S

SYMPOSIUM DIGEST, P43-45 JUNE 1974.

823
TESTING OF ELECTRONIC INDUSTRIES ASSOCIATION LAND-MOBILE

COMMUNICATION ANTENNA GAIN STANDARDS AT THE NATIONAL BUREAU OF

STANDARDS;

TAGGART, H. E.; SHAFER, J. F.

IEEE (INSTITUTE OF ELECTRICAL AND ELECTRONICS ENGINEERS)

TRANS. ON VEHICULAR TECHNOLOGY VT-27, NR P259-264, NOVEMBER 1978.

FIXED AND BASE STATION ANTENNAS;

TREADO, M. J.; TAGGART, H. E.; NELSON, R. E.; WORKMAN, J. L. NILECJ-STD-0204.00, 16P 1977.

THE ORBITING STANDARDS PLATFORM;

DOUGHERTY, H. T.; ESTIN, A. J.; MORGAN, W. L.; WOODRUFF, J. J.

PROCEEDINGS OF THE ANTENNA APPLICATIONS SYMPOSIUM (1978) HELD

AT URBANA-CHAMPAIGN, IL. ON SEPTEMBER 20-22, 1978, Pl-9, 1978.

AN ERROR ANALYSIS FOR THE MEASUREMENT OF SATELLITE EIRP USING A

CALIBRATED RADIO STAR;

DAYWITT WILLIAM C.

JNL. IEEE TRANS. INSTRUM. MEAS. VIM27 N3 P253-258 SEPTEMBER 1978.

FEASIBILITY STUDY OF ORBITING STANDARDS PLATFORM;

ESTIN, A. J.; BAIRD, R. C.

NBSIR-78-886, 47p 1978.

A PRECISION EARTH-TERMINAL SYSTEM FOR ACCURATE C/KT, G/T, AND EIRP

MEASUREMENTS WITH A CALIBRATED RADIO STAR;

DAYWITT, W. C.

PROC. URSI INT. SYMP. ON MEASUREMENTS IN TELECOMMUNICATIONS,

LANNION, FRANCE, OCTOBER 3-7 Pl-4, 1977.

EARTH TERMINAL MEASUREMENT SYSTEM OPERATIONS MANUAL;

WAIT, DAVID F.

NBSIR-78-879, 266p 1978.

ATMOSPHERIC PROPAGATION EQUATIONS USED IN THE NBS EARTH TERMINAL

MEASUREMENT SYSTEM;

DAYWITT, W. C.

NBSIR-78-883, 46p 1978.

LASER FAR-FIELD BEAM-PROFILE MEASUREMENTS BY THE FOCAL PLANE

TECHNIQUE;

DAY, G. W. STUBENRAUCH, C. F. NBS-TN-1001, 52p 1978.

ERROR EQUATIONS USED IN THE NBS EARTH TERMINAL MEASUREMENT SYSTEM;

DAYWITT, WILLIAM C.

NBSIR-78-869, 32p 1977.

THE USE OF THREE TERM RECURSION RELATIONS FOR NUMERICAL COMPUTATIONS AS

APPLIED TO NEAR-FIELD SPHERICAL SCANNING;

LEWIS, RICHARD L.

PROCEEDINGS OF INTERNATIONAL UNION OF RADIO SCIENCE (URSI)

INTERNATIONAL SYMPOSIUM, 20-24 JUNE 1977 P224-226 1977.

824
NEAR-FIELD ANTENNA MEASUREMENTS ON A CYLINDRICAL SURFACE: A SOURCE SCATTERING

MATRIX FORMULATION;

YAGHJIAN, ARTHUR D. NBS-TN-696-REV, 42p 1977.

RESULTS OF SPHERICAL NEAR-FIELD MEASUREMENTS ON NARROW-BEAM ANTENNAS;

NEWELL, ALLEN C; REPJAR, ANDREW.

PROCEEDINGS OF CONFERENCE ON PRECISION ELECTROMAGNETIC

MEASUREMENTS, P382-385 1976.

ADVANTAGES AND DISADVANTAGES OF PLANAR, CIRCULAR CYLINDRICAL, AND

SPHERICAL SCANNING AND DESCRIPTION OF THE NBS ANTENNA SCANNING

FACILITIES;

WACKER PAUL F. NEWELL ALLEN C.

PROCEEDINGS EUROPEAN SPACE AGENCY SYMPOSIUM, NOORDWIJK

(NETHERLANDS), JUNE 6-8, 1977, P115-121 1977.

SYMMETRY ANALYSIS APPLIED TO WAVE THEORY;

WACKER PAUL F. INT'L IEEE/AP-S SYMPOSIUM DIGEST, 4p 1977.

SYMMETRY ANALYSIS APPLIED TO SCATTERING, INVERSE SCATTERING, AND

ANTENNA PATTERNS: MEASUREMENTS, MOMENT METHOD, AND CHARACTERISTIC

MODES;

WACKER PAUL F. PROCEEDINGS OF* INSTITUTE OF ELECTRICAL AND ELECTRONICS

ENGINEERS/ANTENNAS AND PROPAGATION SOCIETY INT. SYMP. P177-180 1977.

RECENT RESULTS WITH SPHERICAL NEAR-FIELD ANTENNA SCANNING AT THE

NATIONAL BUREAU OF STANDARDS;

WACKER PAUL F. PROCEEDINGS EUROPEAN SPACE AGENCY SYMPOSIUM, NOORDWIJK

(NETHERLANDS), JUNE 6-8, 1977, P159-164 1977.

SATELLITE EARTH TERMINAL G/T MEASUREMENTS;

WAIT, DAVID F. MICROWAVE JNL. 20, N4 P49, 51, 58, APRIL 1977.

A DIRECT APPROACH TO THE DERIVATION OF ELECTRIC DYADIC GREEN'S

FUNCTIONS;

YAGHJIAN, ARTHUR D. AP-SYMPOSIUM DIGEST, P 76-156 OCTOBER 1976 and NBS-TN-1000, P70, 1978.

ERROR EQUATIONS USED IN THE NBS PRECISION G/T MEASUREMENT SYSTEM;

DAYWITT, W. C.

NBSIR-76-842, 21p 1976.

THE ACCURATE DETERMINATION OF MILLIMETER-WAVE ANTENNA CHARACTERISTICS BY

DECONVOLUTION AND EXTRAPOLATION TECHNIQUES;

BAIRD, R. C; KERNS, D. M.

PROC. 1974 MILLIMETER-WAVE CONF., 26-28 MARCH 1974, VOL. 2, E 2-1 THRU E 2-12

14p MARCH 1974.

825
STUDY OF ERRORS IN PLANAR NEAR-FIELD MEASUREMENTS;

NEWELL, ALLEN C; YAGHJIAN, ARTHUR D.

PROCEEDINGS OF IEEE INT. ANTENNAS AND PROPAGATION SYMP.

URBANA/CHAMPAIGN, ILL. 2-4 JUNE 75, SESSION 20, P470-473 1975.

IMPROVED POLARIZATION MEASUREMENTS USING A MODIFIED THREE ANTENNA

TECHNIQUE;

NEWELL, ALLEN C.

PROCEEDINGS OF IEEE INT. ANTENNAS AND PROPAGATION SYMP.

URBANA/CHAMPAIGN, ILL. 2-4 JUNE 75, SESSION 15, P337-340 1975.

UPPER-BOUND ERRORS IN FAR-FIELD ANTENNA PARAMETERS DETERMINED FROM PLANAR

NEAR-FIELD MEASUREMENTS. PART 1: ANALYSIS;

YAGHJIAN, ARTHUR D. NBS-TN-667. 1, 113p OCTOBER 1975.

PLANE-WAVE SCATTERING-MATRIX THEORY OF ANTENNAS AND ANTENNA-ANTENNA

INTERACTIONS: FORMULATION AND APPLICATIONS;

KERNS, DAVID M.

NBSIR-75-824, 99p 1975.

NON-PLANAR NEAR-FIELD MEASUREMENTS: SPHERICAL SCANNING;

WACKER, PAUL F.

NBSIR-75-809, 69p 1975.

ANTENNA MEASUREMENTS AT THE NATIONAL BUREAU OF STANDARDS NEAR-FIELD AND

EXTRAPOLATION TECHNIQUES;

WACKER PAUL F.

PROCEEDINGS COLLOQUIUM ON MICROWAVE- COMMUNICATION (5TH)

BUDAPEST, 24-30 JUNE 74, PME-113-114.

NEAR-FIELD ANTENNA MEASUREMENTS USING A SPHERICAL SCAN: EFFICIENT

DATA REDUCTION WITH PROBE CORRECTION;

WACKER PAUL F.

CPEM DIGEST N113 P286-288 (INSTITUTION OF ELECTRICAL ENGINEERS,

LONDON, ENGLAND) 1974.

COMMENT ON CORRECTION OF ERRORS IN AERIAL FAR-FIELD RADIATION-PATTERN

DETERMINATIONS;

KERNS, D. M.

ELECTRONICS LETTERS, VOL. 7 N24 P706, 2 DECEMBER 1971.

ACCURATE MEASUREMENT OF ANTENNA GAIN AND POLARIZATION AT REDUCED

DISTANCES BY AN EXTRAPOLATION TECHNIQUE;

NEWELL, ALLEN C; BAIRD, RAMON C; WACKER, PAUL F.

IEEE, TRANSACTIONS ON ANTENNAS AND PROPAGATION, AP-21 NO.

P418-431 JULY 1973.

DETERMINATION OF BOTH POLARIZATION AND POWER GAIN OF ANTENNAS BY A GENERALIZED

3-ANTENNA MEASUREMENT METHOD;

NEWELL, A. C; KERNS, D. M.

ELECTRONICS LETTERS, VOL. 7 NO. 3 P68-70, JANUARY 1971.

826
SOME RECENT NEAR-FIELD ANTENNA MEASUREMENTS;

STUBENRAUCH, C. F.

PROCEEDINGS 1979 ANTENNA APPLIC. SYMP, SEPTEMBER 1979

SYNTHESIS OF BROADBAND ANTENNA ARRAYS AS POSSIBLE OVER-THE-HORIZON RADARS;

MA, M. T. NBS-MONOGRAPH, JUNE 1979.

PLANE-RADIAL SCANNING TECHNIQUES WITH PROBE CORRECTION; NATURAL

ORTHOGONALITIES WITH RESPECT TO SUMMATION ON PLANAR MEASUREMENT LATTICES;

WACKER, P. F. DIGEST OF THE IEEE/APS SYMPOSIUM, FEBRUARY 1979.

A QUALITATIVE SURVEY OF NEAR-FIELD ANALYSIS AND MEASUREMENT;

WACKER P. F. NBSIR-79-1602, JANUARY 1979.

827
EMI AND RADIATION HAZARDS METROLOGY

picosecond pulse generators using microminiature mercury switches;

Andrews, james r. NBSIR-74-377, 45p 1974.

IMPULSE SPECTRAL INTENSITY — WHAT IS IT; ARTHUR, M. G.

NBSIR-74-365, 21P MAY 1974.

ELECTROMAGNETIC NOISE IN ROBENA NO. 4 COAL MINE;

BENSEMA, W. D.; KANDA, MOTOHISA; , ADAMS, J. W.

NBS-TN-654, 197p APRIL 1974.

AMPLITUDE STATISTICS OF ELECTROMAGNETIC NOISE IN COAL MINES;

KANDA, M.; ADAMS, J. W.

PROCEEDINGS THRU-THE-EARTH ELECTROMAGNETICS WORKSHOP,

COLORADO SCHOOL OF MINES, GOLDEN, COLO. P156-160, 15-17 AUGUST 1973,

AND URSI DIGEST, IEEE TRANS. A&P MAY 1975.

SURVEY REPORT OF THE U.S. BUREAU OF MINES ELECTROMAGNETIC NOISE

MEASUREMENT PROGRAM;

ADAMS, J. W.; TAGGART, H. E. ; SPAULDING, A. D. NBS-10723, 38p 1971.

CHARACTERISTIC IMPEDANCE OF A RECTANGULAR COAXIAL LINE WITH OFFSET INNER

CONDUCTOR;

TIPPET, JOHN C; CHANG, DAVID C.

JNL. IEEE TRANS. MICROWAVE THEORY TECH. VMTT-26 Nil, p876-883

NOVEMBER 1978.

PROCEEDINGS OF THE 1978 ELECTROMAGNETIC INTERFERENCE WORKSHOP;

ARTHUR, M. G.

NBS-SP-551, 61p JULY 1979.

COMPARISON AND SELECTION OF TECHNIQUES FOR MEASURING EM RADIATED

EMISSIONS AND SUSCEPTIBILITY OF LARGE EQUIPMENT;

CRAWFORD, MYRON L. PROCEEDINGS SYMP. AND TECHNICAL EXHIBITION ON ELECTROMAGNETIC

COMPATIBILITY (3RD), HELD AT ROTTERDAM (NETHERLANDS), ON MAY 1-3, 1979.

PAPER IN ELECTROMAGNETIC COMPATIBILITY 1979, P115-122 MAY 79.

USING A TEM CELL FOR EMC MEASUREMENTS OF ELECTRONIC EQUIPMENT;

CRAWFORD, M. L.; WORKMAN, J. L. NBS-TN-1013, 72p APRIL 1979.

VARIATIONAL EXPRESSION FOR THE SCATTERING MATRIX OF A COAXIAL LINE STEP

DISCONTINUITY AND ITS APPLICATION TO AN OVER MODED COAXIAL TEM CELL;

SREENIVASIAH, I.; CHANG, DAVID C.

NBSIR-79-1606, 48p MAY 1979.

CONSTRUCTION OF A LARGE TRANSVERSE ELECTROMAGNETIC CELL;

DECKER, W. F.; CRAWFORD, M. L.; WILSON, W. A.

NBS-TN-1011, 89p FEBRUARY 1979.

828
A NEW APPROXIMATION FOR THE CAPACITANCE OF A RECTANGULAR, COAXIAL, STRIP

TRANSMISSION LINE;

TIPPET, J. C; CHANG, D. C.

IEEE (INSTITUTE OF ELECTRICAL AND ELECTRONICS ENGINEERS)

TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, P602-604, SEPTEMBER 1976.

BROADBAND ISOTROPIC ANTENNA WITH FIBER-OPTIC LINK TO A CONVENTIONAL RECEIVER;

LARSEN, EZRA B.; ANDREWS, JAMES R.; BALDWIN, EUGENE E. PATENT-4 091 327, PATENTED MAY 23, 1978.

EXPANDING THE BANDWIDTH OF TEM CELLS FOR EMC MEASUREMENTS;

CRAWFORD, M. L.; WORKMAN, J. L.; THOMAS, C. L. IEEE TRANSACTIONS ON ELECTRO-MAGNETIC COMPATIBILITY EMC 20 N3

P368-375 AUGUST 1978.

SHF IMPULSE GENERATOR;

ANDREWS, JAMES R. ; BALDWIN, EUGENE E. NBSIR-78-888, 76p JUNE 1978.

UHF IMPULSE GENERATOR;

ANDREWS, JAMES R.; BALDWIN, EUGENE E.

NBSIR-78-880, 30p APRIL 1978.

NEAR-FIELD ELECTRIC FIELD STRENGTH LEVELS OF EM ENVIRONMENTS

APPLICABLE TO AUTOMOTIVE SYSTEMS;

ADAMS, J. W.; KANDA, M. ; SHAFER, J.; WU, Y. PROCEEDINGS OF IEEE INTERNATIONAL SYMPOSIUM ON

ELECTROMAGNETIC COMPATIBILITY, SEATTLE, WASH. 2-4 AUGUST 1977,

PP336-343.

GENERATION OF EM SUSCEPTIBILITY TEST FIELDS USING A LARGE

ABSORBER-LOADED TEM CELL;

CRAWFORD, M. L. WORKMAN, J. L. THOMAS, C. L. IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, IM26 N3

P336-343 SEPTEMBER 1977.

CONVERTING A RECTANGULAR SHIELDED ENCLOSURE INTO A TEM TRANSMISSION CELL FOR

EMI MEASUREMENTS;

CRAWFORD, M. L.; THOMAS, C. L. IEEE 1977 INTERNATIONAL SYMPOSIUM ON EMC (SEATTLE WASH.,) AUGUST 2-5, 1977.

SPECTRUM AMPLITUDE— DEFINITION, GENERATION AND MEASUREMENT;

ANDREWS, JAMES R. ; ARTHUR, M. GERALD

NBS-TN-699. lOOp OCTOBER 1977.

ELECTROMAGNETIC INTERFERENCE (EMI) MEASUREMENTS FOR AUTOMOTIVE

APPLICATIONS;

ADAMS, J. W.; CRAWFORD, M. L.; SHAFER, J. F. PROCEEDINGS OF SAE AUTOMOTIVE CONGRESS CONFERENCE, DETROIT,

MICH. 23-27 FEBRUARY 1976 Pl-6 1976.

829
EXPERIMENTAL EVALUATION OF THE RADIATION CHARACTERISTICS OF DIPOLE SOURCES

ENCLOSED IN A TEM TRANSMISSION CELL;

CRAWFORD, M. L.

PROCEEDINGS OF CONFERENCE ON PRECISION ELECTROMAGNETIC

MEASUREMENTS, P57-59 BOULDER, CO. JUNE 28 - JULY 1, 1976.

AMPLITUDE, TIME, AND FREQUENCY STATISTICS OF QUASI-IMPULSIVE NOISE;

BENSEMA, W. D.

PROCEEDINGS 1977 EMC SYMPOSIUM, MONTREUX (SWITZERLAND), JUNE 18-30, 1977,

P347-352 .

A NOISE SPECTRUM MEASUREMENT SYSTEM USING THE FAST FOURIER TRANSFORM;

BENSEMA, W. D.

IEEE TRANSACTIONS ON ELECTRO-MAGNETIC COMPATIBILITY EMC-19,

N2 P37-43 MAY 77.

SOME RECENT DEVELOPMENTS IN THE CHARACTERIZATION AND MEASUREMENT OF HAZARDOUS

ELECTROMAGNETIC FIELDS;

BOWMAN, RONALD R. PROCEEDINGS OF INTERNATIONAL SYMPOSIUM BIOLOGIC EFFECTS AND

HEALTH HAZARDS OF MICROWAVE RADIATION, OCTOBER 15-18, 1973, WARSAW (POLAND),

P217-227.

NONMETALLIC ELECTRODE SYSTEM FOR RECORDING EEG AND ECG IN ELECTROMAGNETIC FIELDS;

FLANIGAN, W. F. JR.; BOWMAN, R. R.; LOWELL, W. R. PHYSIOLOGY AND BEHAVIOR, VOL. 18 NO. 3 P531-533 1977.

TECHNIQUES FOR MEASUREMENT OF ELECTROMAGNETIC RADIATION AND

SUSCEPTIBILITY OF ELECTRONIC EQUIPMENT;

CRAWFORD, MYRON L. PROCEEDINGS OF SYMP. AND TECHNICAL EXHIBITION ON ELECTROMAGNETIC COMPATIBILITY

(1ST), MONTREUX, SWITZERLAND, MAY 20-22, 1975, P38-44.

QUANTIFYING HAZARDOUS ELECTROMAGNETIC MICROWAVE FIELDS: PRACTICAL

CONSIDERATIONS;

BOWMAN, RONALD R. PROCEEDINGS OF BIOLOGICAL EFFECTS AND HEALTH IMPLICATIONS OF MICROWAVE

RADIATION, HELD AT RICHMOND, VA. ON SEPTEMBER 17-19, 1969,

BRH/DBE 70-2, P204-209, U.S. DEPT. OF HEALTH, EDUCATION, AND WELFARE,

ROCKVILLE, MD. JUNE 70.

SENSITIVE ISOTROPIC ANTENNA WITH FIBER-OPTIC LINK TO A CONVENTIONAL RECEIVER;

LARSEN, E. B.; ANDREWS, J. R.; BALDWIN, E. E. NBSIR-75-819. 114p SEPTEMBER 1976.

AN ANALYTICAL AND EXPERIMENTAL DETERMINATION OF THE CUTOFF

FREQUENCIES OF HIGHER-ORDER TE MODES IN A TEM CELL;

TIPPET, JOHN C; CHANG, DAVID C; CRAWFORD, MYRON L.

NBSIR-76-841, 29p JUNE 1976.

CALIBRATION TECHNIQUES FOR ELECTROMAGNETIC HAZARD METERS: 500 MHZ TO 20 GHz;

BOWMAN, RONALD R. NBSIR-75-805, 38p 1976.

830
A PROBE FOR MEASURING TEMPERATURE IN RADIO-FREQUENCY-HEATED MATERIAL;

BOWMAN, RONALD R. IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VMTT-24

Nl P43-45 JANUARY 1976.

EXPERIMENTAL RF FIELD POLARIZATION METER;

FITZGERRELL, R. G.

FAA-RD-74-188, 41p 1975.

EVALUATION OF REFLECTIVITY LEVEL OF ANECHOIC CHAMBERS USING

ISOTROPIC, 3-DIMENSIONAL PROBING;

CRAWFORD, M. L. PROCEEDINGS OF INTERNATIONAL IEEE/AP-S SYMPOSIUM DIGEST, ATLANTA, GA.

10-12 JUNE 1974, P28-34.

GENERATION OF STANDARD EM FIELDS FOR CALIBRATION OF POWER DENSITY

METERS 20 kHZ TO 1000 MHz;

CRAWFORD, M. L. NBSIR-75-804, 45p JANUARY 1975.

ELECTROMAGNETIC NOISE IN LUCKY FRIDAY MINE;

SCOTT, W. W.; ADAMS, J. W. ; BENSEMA, W. D.; DOBROSKI, H. NBSIR-74-391, 139p 1974.

ELECTROMAGNETIC NOISE IN MCELROY MINE;

KANDA, MOTOHISA; ADAMS, JOHN W.; BENSEMA, WILLIAM D.

NBSIR-74-389, 172p JUNE 1974.

GENERATION OF STANDARD EM FIELDS USING TEM TRANSMISSION CELLS;

CRAWFORD, M. L. IEEE TRANSACTIONS ON ELECTRO-MAGNETIC COMPATIBILITY, VOL. EMC 16 N4 P189-195

NOVEMBER 1974.

ELECTROMAGNETIC NOISE IN ITMANN MINE;

BENSEMA, WILLIAM D.; KANDA, MOTOHISA.; ADAMS, JOHN W.

NBSIR-74-390, 114p JUNE 1974.

SURFACE MAGNETIC FIELD NOISE MEASUREMENTS AT GENEVA MINE;

ADAMS, JOHN W.; BENSEMA, WILLIAM D.; TOMOEDA, N. C.

NBSIR-74-369, 41p JUNE 1974.

ELECTROMAGNETIC NOISE IN GRACE MINE;

ADAMS, JOHN W.; BENSEMA, WILLIAM D.; KANDA, MOTOHISA.

NBSIR-74-388, 139p JUNE 1974.

VARIABLE IMPEDANCE COAXIAL DEVICE WITH RELATIVE ROTATION BETWEEN CONDUCTORS;

CRUZ, JOSE E. PAT-APPL-487 099, PATENT-3 417 350, 4p 1965.

SIMULATOR FOR ATMOSPHERIC RADIO NOISE;

BENSEMA, WILLIAM D.; COON, ROBERT M.; BERRY, WESLEY M. ; WATTERSON, CLARK C;

BOLTON, EARL C.

PAT-APPL-44 842, PATENT-3 617 925, 8P 1970.

831
COAL MINE MAGNETIC-FIELD NOISE MEASUREMENTS;

BENSEMA, W. D.

PROCEEDINGS OF WEST VIRGINIA CONFERENCE ON COAL MINE ELECTROTECHNOLOGY (1ST),

WEST VIRGINIA UNIV., MORGANTOWN, 2-4 AUGUST 1972 PXII-1— XI 1-12, 12p 1972.

ANALYTICAL AND NUMERICAL TECHNIQUES FOR ANALYZING AN ELECTRICALLY

SHORT DIPOLE WITH A NONLINEAR LOAD;

KANDA, MOTOHISA.

NBSIR-78-898, 21p 1978.

A RELATIVELY SHORT CYLINDRICAL BROADBAND ANTENNA WITH TAPERED

RESISTIVE LOADING FOR PICOSECOND PULSE MEASUREMENTS;

KANDA, MOTOHISA.

NBSIR-77-861. 47 P. AUGUST 1977 AND IEEE TRANS. ANTENNAS PROPAG. AP-26 N3

P439-447 MAY 1978.

THE CHARACTERISTICS OF A RELATIVELY SHORT BROADBAND LINEAR ANTENNA WITH

TAPERED RESISTIVE LOADING;

KANDA, M.

PROC. INT. SYMP. ON ANTENNAS AND PROPAGATION SOCIETY, JUNE 20-22, 1977. IEEE

ANTENNAS AND PROPAGATION SOCIETY DIGEST, P230-233, JUNE 20-22, 1977.

A BROADBAND ANTENNA WITH TAPERED RESISTIVE LOADING FOR EMI MEASUREMENTS;

KANDA, MOTOHISA.

PROCEEDINGS IEEE INT. SYMP. ON ELECTROMAGNETIC COMPATIBILITY,

HELD AT SEATTLE, WASHINGTON, ON AUGUST 2-4, 1977, P13-18.

TRANSIENTS IN RESISTI VELY LOADED ANTENNAS AND THEIR COMPARISON WITH CONICAL

ANTENNAS AND TEM HORNS;

KANDA, MOTOHISA.

NBSIR-78-876, 38p MARCH 1978.

SURVEYS OF ELECTROMAGNETIC FIELD INTENSITIES NEAR REPRESENTATIVE

HIGHER-POWER FAA TRANSMITTING ANTENNAS;

LARSEN, EZRA B.; SHAFER, JOHN F.

FAA-RD-77 179 P DECEMBER 1977.

DESIGN OF A VAN-TOP LOW-PROFILE HF ANTENNA;

MA, M. T.; FITZGERRELL, R. G.

OTR-77-131, 47p OCTOBER 1977.

THE CHARACTERISTICS OF BROADBAND, ISOTROPIC ELECTRICAL FIELD AND

MAGNETIC FIELD PROBES;

KANDA, MOTOHISA.

NBSIR-77-868, 35p NOVEMBER 1977.

MEASUREMENT OF RF POWER-ABSORPTION IN BIOLOGICAL SPECIMENS (10 TO 100 MHZ);

GREENE, FRANK M.

NBS-TN-687, 33p NOVEMBER 1976.

832
USING FIBER OPTICS IN A BROADBAND, SENSITIVE, ISOTROPIC ANTENNA - 15 KHZ TO

150 MHZ;

LARSEN, E. B.; ANDREWS, J. R. PROCEEDINGS OF IEEE INTERNATIONAL SYMPOSIUM ON ELECTROMAGNTIC COMPATIBILITY,

7p WASH. D.C. JULY 13-15, 1976.

COMPLETION OF THE PROGRAM TO EVALUATE/IMPROVE INSTRUMENTATION AND

TEST METHODS FOR ELECTROEXPLOSIVE DEVICE SAFETY QUALIFICATION;

HUDSON, PAUL A.; MELQUIST, DEAN G. ; ONDREJKA, ARTHUR R.; WERNER, P.E.

NBSIR-73-323 37p 1973, and NBSIR-74-379, 37p 1974.

CALIBRATION OF IMPULSE NOISE GENERATORS;

REEVE, GEROME R. NBSIR-73-343, 78p OCTOBER 1973.

DEVELOPMENT OF ELECTRIC AND MAGNETIC NEAR-FIELD PROBES;

GREENE, FRANK M.

NBS-TN-658, 55p JANUARY 1975.

ASYMMETRIC VERSUS SYMMETRIC TEM CELLS FOR EMI MEASUREMENTS;

CRAWFORD, M. L.; WORKMAN, J. L. PROCEEDINGS IEEE INT. SYMP. ON ELECTROMAGNETIC COMPATIBILITY,

HELD AT ATLANTA, GA. ON JUNE 20-22, 1978, P204-210 1979.

HIGHER ORDER MODES IN RECTANGULAR COAXIAL LINE WITH INFINITELY THIN INNER

CONDUCTOR;

TIPPET, JOHN C; CHANG, DAVID C.

NBSIR-78-873, 41p MARCH 1978.

RADIATION CHARACTERISTICS OF DIPOLE SOURCES LOCATED INSIDE A

RECTANGULAR, COAXIAL TRANSMISSION LINE;

TIPPET, JOHN C; CHANG, DAVID C.

NBSIR-75-829, 35p JANUARY 1976.

BROADBAND PULSED/CW CALIBRATION SIGNAL STANDARD FOR FIELD INTENSITY METER

(FIM) RECEIVERS;

SIMPSON, PHILIP A.

NBSIR-74-371. 53p JUNE 1974.

DEVELOPMENT OF NEAR-FIELD ELECTRIC ENERGY DENSITY METER MODEL EDM-2;

BELSHER, DONALD R. DEPARTMENT OF HEALTH, EDUCATION, AND WELFARE PUBL. NO. (NIOSH) 75-140, 26P

MARCH 1975.

QUANTIFYING HAZARDOUS MICROWAVE FIELDS: ANALYSIS;

WACKER PAUL F. PROCEEDINGS OF SYMPOSIUM ON BIOLOGICAL EFFECTS AND HEALTH

IMPLICATIONS OF MICROWAVE RADIATION, HELD AT RICHMOND, VIRGINIA ON

SEPTEMBER 17-19, 1969, BRH/DBE 70-2, P197-203, U.S. DEPARTMENT OF HEALTH,

EDUCATION, AND WELFARE, ROCKVILLE, MD. JUNE 1970.

833
QUANTIFYING HAZARDOUS ELECTROMAGNETIC FIELDS: SCIENTIFIC BASIS AND PRACTICAL

CONSIDERATIONS;

WACKER, PAUL F.; BOWMAN, RONALD R. IEEE TRANS. MICROWAVE THEORY AND TECHNIQUES, VOL. MTT-19 NO. 22 FEBRUARY 1971.

METHODS OF CALIBRATING MICROWAVE HAZARD METERS;

BAIRD, R. C.

PAPER GIVEN AT INTL

1 SYMPOSIUM WARSAW OCTOBER 15-18, 1973 AND PUBLISHED

"BIOLOGICAL EFFECTS & HEALTH HAZARDS OF MICROWAVE RADIATION" PROCEEDINGS OF AN

INTL' SYMPOSIUM WARSAW OCTOBER 15-18, 228-236p OCTOBER 1973.

IMPROVED TECHNIQUES AND INSTRUMENTATION FOR EMC MEASUREMENTS;

CRAWFORD, MYRON L. IEEE 1976 INTERNATIONAL SYMPOSIUM ON ELECTROMAGNETIC COMPATIBILITY (WASH.,

D.C.) 7p JULY 13-15 1976.

TIME AND AMPLITUDE STATISTICS FOR ELECTROMAGNETIC NOISE IN MINES;

KANDA, MOTOHISA.

NBSIR-74-378, 60p 1974.

DEVELOPMENT AND CONSTRUCTION OF AN ELECTROMAGNETIC NEAR-FIELD

SYNTHESIZER;

GREENE, FRANK M.

NBS-TN-652, 46p MAY 1974.

CALIBRATION OF RADIO RECEIVERS TO MEASURE BROADBAND INTERFERENCE;

LARSEN, EZRA B.

NBSIR-73-335, 76p 1973.

EMI - A PROBLEM OF GROWING CONCERN

REPORT OF THE 63RD NATIONAL CONFERENCE ON WEIGHTS & MEASURES, 1978;

MILLER, C. K. S.

NBS-SP-532, 12p, 1978

A BROADBAND ISOTROPIC, REAL-TIME, ELECTRIC FIELD SENSOR 9BIRES0 USING

RESISTIVELY LOADED DIPOLES;

KANDA, M. ; RIES, F. X.; BELSHER, D. R.

IEEE TRANS. EMC & NBSIR 79-1622, SEPTEMBER 1979.

DISPERSION AND ATTENUATION CHARACTERISTICS OF MODES IN A TEM-CELL WITH A LOSSY

DIELECTRIC SLAB;

TIPPET, J. C; CHANG, C.

NBSIR 79-1615, JULY, 1979.

THE EFFECTS OF RESISTIVE LOADING ON TEM HORNS

KANDA, M.

NBSIR-79-1601 AND IEE TRANS. ON ANTENNAS & PROPAGATION;

30 P, JANUARY 1979.

THE CHARACTERISTICS OF A TRAVELING-WAVE, LINEAR ANTENNA WITH A NONLINEAR LOAD;

KANDA, M.

DIGEST ABSTRACT, 1979 IEEE APIS SYMPOSIUM, JANUARY 1979.

CHARACTERISTICS OF A CISPR/VDE FAR-FIELD EMI TEST SITE WITH GROUND SCREEN

TAGGART, H. E.;

PROCEEDINGS OF THE THIRD ROTTERDAM EMC SYMPOSIUM

DECEMBER 1978.

834
ELECTROMAGNETIC PROPERTIES OF MATERIALS

ELECTROMAGNETIC REMOTE SENSING OF INHOMOGENEOUS MEDIA;

BEREUTER, WOLFGANG A.; CHANG, DAVID C.

NBSIR-76-851, 21p 1977.

MICROWAVE MEASUREMENT OF COAL LAYER THICKNESS;

ELLERBRUCH, DOYLE A.; ADAMS, JOHN W.

NBSIR-74-387, 33p 1974.

CONTINUOUS LIQUID LEVEL MEASUREMENTS WITH TIME-DOMAIN REFLECTOMETRY

;

CRUZ, J. E.; ROGERS, E. H.; HIESTER, A. E. ADVANCES IN CRYOGENIC ENGINEERING, V18 P323-327, 1973. PAPER H-4.

HYDROGEN SLUSH DENSITY REFERENCE SYSTEM;

WEITZEL, D. H.; LOWE, L. T. ; ELLERBRUCH, D. A.; CRUZ, J. E.; SINDT, C. F. NASA-CR-124764, AND NBS-9796, 83p 1971.

INSTRUMENTATION FOR HYDROGEN SLUSH STORAGE CONTAINERS;

COLLIER, R. S.; CRUZ, J. E.; ELLERBRUCH, D. A.; LOWE, L. T.; WEITZEL, D, H.

NASA-CR-119241, AND NBS-9793, 99p 1971.

FEASIBILITY STUDY ON THE USE OF A MICROWAVE SYSTEM FOR THE

NONDESTRUCTIVE EVALUATION OF HISTORIC ADOBE STRUCTURES;

BELSHER, D. R. NBSIR-79-1610, 33p 1979.

ELECTROMAGNETIC BOUNDARY-VALUE PROBLEMS BASED UPON A MODIFICATION OF RESIDUE

CALCULUS AND FUNCTION THEORETIC TECHNIQUES;

MONTGOMERY, JAMES PATRICK; CHANG, DAVID C.

NBS/MN-164, 126p 1979.

ELECTROMAGNETIC SCATTERING PROPERTIES OF SOILS AND SNOW;

ELLERBRUCH, D. A.; JESCH, R. L.; JONES, R. N. ; BUSSEY, H. E.; BOYNE, H. S.

PROCEEDINGS INT. SYMP. ON REMOTE SENSING OF ENVIRONMENT

(12TH), HELD AT ANN ARBOR, MI ON 20-26 APRIL 1978, V2 P957-974.

PRELIMINARY RESULTS OF PASSIVE MICROWAVE SNOW EXPERIMENT DURING

FEBRUARY AND MARCH 1978;

CHANG, A. T. C; SHIUE, J. C; BOYNE, H.; ELLERBRUCH, D. ; COUNAS, G.

NASA-TP-1408, REPT-7802-F22, 114p 1979.

DIELECTRIC MEASUREMENTS OF FIVE DIFFERENT SOIL TEXTURAL TYPES AS

FUNCTIONS OF FREQUENCY AND MOISTURE CONTENT;

JESCH, RAMON L. NBSIR-78-896, 27p 1978.

ELECTRICAL CHARACTERISTICS OF CORN, WHEAT, AND SOYA IN THE 1-200 MHZ RANGE;

JONES, R. N.; BUSSEY, H. E.; LITTLE, W. E.; METZKER, R. F.

NBSIR-78-897, 70p 1978.

ELECTROMAGNETIC TECHNIQUE OF MEASURING COAL LAYER THICKNESS;

ELLERBRUCH, DOYLE A.; BELSHER, DONALD R. IEEE TRANSACTIONS ON GEOSCIENCE ELECTRONICS, GE16 N2 P126-133

APRIL, 1978.

835
MICROWAVE CHARACTERISTICS OF SNOW;

ELLERBRUCH, D. A.; LITTLE, W. E.; BOYNE, H. S.; BACHMAN, D. D

PROC. WESTERN SNOW CONF. ALBUQUERQUE, N. MEX. 7P, APRIL 18-21, 1977.

FM-CW ELECTROMAGNETIC TECHNIQUE OF MEASURING COAL LAYER THICKNESS;

ELLERBRUCH, DOYLE A.; BELSHER, DONALD R. NBSIR-76-840, 38p 1976.

ELECTROMAGNETIC ATTENUATION PROPERTIES OF CLAY AND GRAVEL SOILS;

ELLERBRUCH, DOYLE A.

NBSIR-74-381, 25p 1974.

APPLICATION OF MEASUREMENTS OF ELECTROMAGNETIC PHENOMENA TO

OCEANOGRAPHY;

ELLERBRUCH, DOYLE A.

PROCEEDINGS OF UJNR/MEC SYMPOSIUM (1ST), RECORD OF THE U.S.

-JAPAN UJNR JOINT SYMPOSIUM ON MARINE ELECTRONICS, TOKYO, JAPAN,

9 OCTOBER 1972, P23-30. (JAPANESE ELECTRICAL INDUSTRIAL COMMITTEE, TOYKO,

JAPAN).

OPEN CIRCUITED COAXIAL RESONATOR FOR HIGH SENSITIVITY DIELECTRIC

MEASUREMENTS, APPLICATION TO LUNAR SOIL 70051-20;

BUSSEY, HOWARD E. PROCEEDINGS LUNAR AND PLANETARY SCIENCE CONF. (10TH), HELD AT

NATIONAL AERONAUTICS AND SPACE ADMINISTRATION, HOUSTON, TX. LYNDON B. JOHNSON

SPACE CENTER ON MAR 19-23, 1979. ABSTRACT IN LUNAR AND PLANETARY SCIENCE, PT1

P169-171, 1979.

DIELECTRIC MEASUREMENTS OF LUNAR SOIL;

BUSSEY, HOWARD E. PROCEEDINGS LUNAR AND PLANETARY SCIENCE CONF. (10TH), HELD AT

NATIONAL AERONAUTICS AND SPACE ADMINISTRATION, HOUSTON, TX. LYNDON B. JOHNSON

SPACE CENTER ON MARCH 19-23, 1979. ABSTRACT IN LUNAR AND PLANETARY SCIENCE,

PT1 P140-142, 1979 (LUNAR AND PLANETARY INST. HOUSTON, TX.).

SCATTERING BY A LOSSY DIELECTRIC CIRCULAR CYLINDRICAL MULTILAYER,

NUMERICAL VALUES;

BUSSEY, HOWARD E.; RICHMOND, JACK H. IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, V23 N5

P723-725 SEPTEMBER 1975.

RAPPORT SUR LA COMPARAISON INTERNATIONALE DES MESURES DE PERMITTIVITE COMPLEXE

A 9 GHZ (INTERNATIONAL COMPARISON OF COMPLEX PERMITTIVITY MEASUREMENT AT 9

GHZ);

BUSSEY, H. E. PROCEEDINGS OF COMITE CONSULTATIF, D'ELECTRICITE COMITE

INTERNATIONAL DES POIDS ET MESURES, SEVRES, FRANCE, 12-13 OCTOBER 1972,

P124-137 1972.

INTERNATIONAL COMPARISON OF COMPLEX PERMITTIVITY MEASUREMENT AT 9 GHZ;

BUSSEY, HOWARD E.; MORRIS, DEREK; ZALTSMAN, E. B.

IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VIM23

N3 P235-239 SEPTEMBER 1974.

836
BURIED ANTENNA PERFORMANCE: DEVELOPMENT OF SMALL RESONANT BURIED

ANTENNAS;

BUSSEY, HOWARD E.; LARSEN, EZRA B. RADC-TR-74-169M FINAL REPORT, AD 783274, 104p JUNE 1974.

TEST RESULTS FOR THE MOORING LINE DATA LINE;

ELLERBRUCH, DOYLE A.

NBSIR-73-341, 46p 1973.

RF (RADIOFREQUENCY) TOTAL MASS GAUGING IN LARGE STORAGE CONTAINERS: EMPTY

TANK MODES;

COLLIER, R. S.; ELLERBRUCH, DOYLE.

NBSIR-73-346, 29p 1973.

MASS QUANTITY GAUGING BY RF MODE ANALYSIS;

COLLIER, R. S.; ELLERBRUCH, D. ; CRUZ, J. E.; STOKES, R. W.; LUFT, P. E.

NASA-CR-133393, AND NBSIR-73-318, 184p 1973.

MICROWAVE METHODS FOR CRYOGENIC LIQUID AND SLUSH INSTRUMENTATION;

ELLERBRUCH, D. A.

IN IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT- IM-19,

N4 P412-416 NOVEMBER 1970.

WAVELENGTH OF A SLOTTED RECTANGULAR LINE CONTAINING TWO DIELECTRICS;

BUSSEY, HOWARD E. NBSIR-73-326, 18p 1973.

MICROWAVE MEASUREMENTS OF SNOW STRATIGRAPHY AND WATER EQUIVALENCE;

BOYNE, H. S. AND ELLERBRUCH, D. A.

PROCEEDINGS OF THE WESTERN SNOW CONFERENCE, APRIL 1979

A COMPARISON OF CENTRIFUGE AND FREEZING CALORIMETER METHODS FOR MEASURING FREE

WATER IN SNOW;

JONES, R. N.

NBSIR 79-1604, APRIL 1979.

HIGH RESOLUTION SENSING TECHNIQUES FOR SLOPE STABILITY STUDIES;

JESCH, R. L.; JOHNSON, R. B.; BELSHER, D. R. ; YAGHJIAN, A. D.; STEPPE, M. C.

AND FLEMING R. W.

FEDERAL HIGHWAY ADMINISTRATION REPORT, FHWA-7-3-0001 , MARCH 1979.

837
MISCELLANEOUS RELEVANT PUBLICATIONS

MICROWAVE ATTENUATION MEASUREMENT SYSTEM (SERIES SUBSTITUTION);

LARSON, WILBUR.; CAMPBELL, EUGENE.

NBS-TN-647, 29p FEBRUARY 1974.

INVESTIGATION OF EFFECT OF ANTENNA SITE AND ANTENNA TYPE ON LF

NON-DIRECTIONAL BEACON PERFORMANCE;

BERRY, LESLIE A.; FITZGERRELL, R. G. ; VOGLER, L. E. FAA-RD-73-174, 87p 1973.

RADAR ABSORBER MEASUREMENT TECHNIQUES AT FREQUENCIES ABOVE 20 GHZ;

NAHMAN, N. S.; ALLRED, C. M. ; ANDREWS, J. R.; HOER, C. A.; LAWTON, R. A.

NBSIR-79-1613, 71p AUGUST 1979.

NBS RF VOLTAGE COMPARATOR;

DRIVER, L. D. ; RIES, F. X.; REBULDELA, G.

NBSIR-78-871, 28p DECEMBER 1978.

AUTOMATIC NETWORK MEASUREMENTS IN THE TIME DOMAIN;

ANDREWS, JAMES R.

PROCEEDINGS OF IEEE, V66 N4 P414-423 APRIL 1978.

TUNNEL DIODE PULSE GENERATOR;

ANDREWS, JAMES R. PAT-APPL-540 471, PATENT-3 967 140, 8P 1975.

IMPULSE GENERATOR SPECTRUM AMPLITUDE MEASUREMENT TECHNIQUES;

ANDREWS, JAMES R. IEEE TRANS. INSTRUM. MEAS. IM-25, N4 P380-384 DECEMBER 1976.

NEW COAXIAL THERMISTOR MOUNTS FOR USE AS PRECISION TRANSFER

STANDARDS;

CRAWFORD, M. L.; SMART, G. R. PROCEEDINGS OF INSTRUMENT SOCIETY OF AMERICA CONFERENCE, PHILADELPHIA,

PENNSYLVANIA, OCTOBER 26-29, 1970, N708-70 Pl-6 1970.

A STANDARD FOR RF MODULATION FACTORS;

REEVE, G. R. ; ARTHUR, M. G.

NBS-TN 1016, 92p SEPTEMBER 1979.

BASEBAND IMPULSE GENERATOR USEFUL TO 5 GHZ;

ANDREWS, JAMES R. ; BALDWIN, EUGENE E. PROCEEDINGS OF 1975 IEEE INTERNATIONAL SYMPOSIUM ON

ELECTROMAGNETIC COMPATIBILITY, SAN ANTONIO, TEX. 7-9 OCTOBER 1975,

Pl-4.

TIME DOMAIN AUTOMATIC NETWORK ANALYZER FOR MEASUREMENT OF RF AND

MICROWAVE COMPONENTS;

GANS, WILLIAM L.; ANDREWS, JAMES R. NBS-TN-672, 177p 1975.

A METHOD FOR DESIGNING MULTI-SCREW WAVEGUIDE TUNERS;

WEIDMAN, M. P.; CAMPBELL, E.

NBS-TN-393, 24p OCTOBER 1970.

838
SELF-MODULATING TRANSDUCERS;

BOWMAN, RONALD.

E0S-7039-FINAL 93p 1968.

MILLIMETER WAVE METROLOGY CAPABILITIES AT NBS;

MILLER, C. K. S.

PUB. IN PROCEEDINGS OF THE 1974 MILLIMETER WAVES TECHNIQUES

CONFERENCE HELD AT SAN DIEGO, CA. ON MARCH 26-28, 1974

IIP.

STANDARDS FOR THE MEASUREMENT OF IMPULSIVE FIELDS RADIATED BY A TEM HORN

ANTENNA;

LAWTON, ROBERT A. ONDREJKA, ARTHUR R. PROCEEDINGS OF UNION RADIO SCIENTIFIQUE INTERNATIONALE COMM.

A SYMPOSIUM HELD AT LANNION, FRANCE ON OCTOBER 3-7, 1977. Pl-4, DECEMBER 1977.

AN INTERNATIONAL INTERCOMPARISON OF VOLTAGE STANDARDS AT 1 GHZ IN COAXIAL TRANSMISSION LINE;

RIES, FRANCIS X. IEEE TRANS. ON INSTRU. & MEAS., VOL. IM-25, NO. 4p SEPTEMBER 1976.

SELF-BALANCING DC-SUBSTITUTION MEASURING SYSTEM;

LARSEN, NEIL T. ; REEVE, GEROME R. PAT-APPL-587 565, 22P 1975.

SELF-BALANCING, D.C. -SUBSTITUTION MEASURING SYSTEM;

LARSEN, NEIL T.; REEVE, GEROME R. PAT-APPL-476 646, 17P 1974.

HOLLOW-CYLINDER WAVEGUIDE ISOLATORS FOR USE AT MILLIMETER WAVELENGTHS;

KANDA, M. ; MAY, W. G. NASA-CR-140760, 21p NOVEMBER 1974.

REFLECTION BEAM ISOLATOR FOR SUBMILLIMETER WAVELENGTHS.

KANDA, M. ; MAY, W. G. NASA-CR-140761, 2p JUNE 1975.

GUIDELINES FOR SPECIFICATION AND PROCUREMENT OF MEASUREMENT

INSTRUMENTATION;

KING, JEANNETTE; PEISER, H. STEFFEN; SANGSTER, RAYMOND C.

NBSIR-78-891, 28p 1978.

FM REPEATER SYSTEMS;

JICKLING, R. M. ; SHAFER, J. F.

NILECJ-STD-0213.00, 16P NOVEMBER 1977.

COLLECTED EXECUTIVE SUMMARIES: STUDIES OF THE NATIONAL MEASUREMENT SYSTEM

1972-75;

SANGSTER, RAYMOND C.

NBSIR-75-947-REV. S., 53p 1977.

839
RELEVANCY (f MEASUREMENTS BY A SYSTEMS APPROACH;

SANGSTER, RAYMOND C.

PROCEEDINGS OF JOINT MEASUREMENT CONFERENCE (1972), BOULDER,

COLORADO, JUNE 21-23, 1972, P31-37 1972.

STUDY OF THE NATIONAL MEASUREMENT SYSTEM. FINAL SUMMARY REPORT,

1972-75;

SANGSTER, RAYMOND C.

NBSIR-75-925, 49p 1976.

TRANSACTIONS MATRIX DESCRIPTION OF THE NATIONAL SYSTEM OF PHYSICAL

MEASUREMENTS;

SANGSTER, RAYMOND C.

NBSIR-75-943, 90p 1976.

COLLECTED EXECUTIVE SUMMARIES: STUDIES OF THE NATIONAL MEASUREMENT SYSTEM

1972-75;

SANGSTER, RAYMOND C.

NBSIR-75-947, 54p 1976.

IMPLEMENTATION OF THE NOTCH TECHNIQUE AS AN RF PEAK PULSE POWER

STANDARD;

SIMPSON, PHILIP A.; HUDSON, PAUL A.

NBS-TN-682, 33p 1976.

REPEATERS FOR LAW ENFORCEMENT COMMUNICATION SYSTEMS-LAW ENFORCEMENT STANDARDS

PROGRAM;

JICKLING, R. M. SHAFER, J. F. LESP-RPT-0206.00, 21p OCTOBER 1974.

A NEW COAXIAL RF-DC AMMETER;

SCOTT W. W. JR.

PROCEEDINGS* OF DIGEST ON PRECISION ELECTROMAGNETIC

MEASUREMENTS CONFERENCE, BOULDER, COLO. JUNE 2-5, 1970, SPECIAL ISSUE CPEM

DIGEST P13.

EVALUATION OF LOW-LOSS/LOW-REFLECTION TWO-PORT DEVICES OR ADAPTERS BY

AUTOMATED MEASUREMENT TECHNIQUES;

JESCH, RAMON L. NBSIR-78-870, 16p 1977.

ULTRASONIC CALORIMETER FOR BEAM POWER MEASUREMENTS FROM 1 TO 15

MEGAHERTZ;

ZAPF, THOMAS L.; HARVEY, MORRIS E.; LARSEN, NEIL T. ; STOLTENBERG, ROBERT E. PROCEEDINGS OF ULTRASONICS SYMPOSIUM (1976), ANNAPOLIS, MD.

29 SEPTEMBER - 1 OCTOBER 1976, IEEE CAT76 CH1120-5SU P573-576 1977.

CHARACTERIZATION OF A H.F. PROBE FOR INTEGRATED CIRCUIT'S

MEASUREMENTS;

JESCH, R. L.; BAILEY, R. A.; TAUSCH, H. J.

PROCEEDINGS, GOVERNMENT MICRO-CIRCUIT APPLICATIONS

CONFERENCE, BOULDER, COLO. 25-27 JUNE 1974 P120-121 1974.

840
REPEATABILITY OF SMA COAXIAL CONNECTORS;

JESCH, R. L. IEEE TRANS. INSTRUM. MEAS. IM-25, N4 P314-320 DECEMBER 1976.

A MICROWAVE VECTOR VOLTMETER SYSTEM;

ROE, KEITH C; HOER, CLETUS A.

NBSIR-76-844. 70p AUGUST 1976.

LASER POWER AND ENERGY MEASUREMENTS;

BOYNE, H. S.

SPIE VOL, 6, 1975.

ULTRASONIC CALORIMETER FOR BEAM POWER MEASUREMENTS;

ZAPF, THOMAS L.; HARVEY, MORRIS E.; LARSEN, NEIL T.; STOLTENBERG,

ROBERT E. NBS-TN-686, 39p SEPTEMBER 1976.

REFLECTION COEFFICIENT STANDARDS FOR AUTOMATED NETWORK ANALYZERS;

LITTLE, W. E. ; YATES, B. C.

CPEM DIGEST 3p 1976.

MEASURING AND MINIMIZING DIODE DETECTOR NONLINEARITY;

HOER, C. A.; ROE, K. C; ALLRED, C. M.

CPEM DIGEST 3p 1976.

USING AN ARBITRARY SIX-PORT JUNCTION TO MEASURE COMPLEX VOLTAGE

RATIOS;

HOER, CLETUS A.; ROE, KEITH C.

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VMTT-23

N12 P978-984 DECEMBER 1975.

CHARACTERIZATION OF A HIGH FREQUENCY PROBE ASSEMBLY FOR INTEGRATED CIRCUIT

MEASUREMENTS;

JESCH, R. L.; HOER, C. A.

NBS-TN-663, 60p 1975.

SEMICONDUCTOR MEASUREMENT TECHNOLOGY: MEASUREMENT OF TRANSISTOR

SCATTERING PARAMETERS;

ROGERS, GEORGE J.; SAWYER, DAVID E.; JESCH, RAMON L. NBS-SPECIAL PUB-400-5, 57p 1975.

AUTOMATED COMPUTER CONTROLLED MEASUREMENTS;

LITTLE, WILLIAM E. PROCEEDINGS OF CONFERENCE ON PRECISION ELECTROMAGNETIC

MEASUREMENTS (1974), LONDON, ENGLAND, 1-5 JULY 74, ABSTRACT IN CPEM DIGEST

N113 P331-332 (INSTITUTION OF ELECTRICAL ENGINEERS, LONDON, ENGLAND) 1974.

A DIRECT APPROACH TO THE DERIVATION OF ELECTRIC DYADIC GREEN'S

FUNCTIONS;

YAGHJIAN, ARTHUR D. NBS-TN-IOOO, 70p 1978.

841
SCATTERING PARAMETERS OF SMA COAXIAL CONNECTOR PAIRS;

ESTIN, A. J.

IEEE TRANS. INSTRUM. MEAS. IM-25, N4 P329-334 DECEMBER 1976.

AN NBS DEVELOPED AUTOMATIC NETWORK ANALYZER;

LITTLE, W.; WAKEFIELD, J.; HEIM, L.; ALLRED, C; ZAPF, T. CPEM DIGEST 3p 1976.

A CALCULABLE MICROWAVE ATTENUATION STANDARD: THE HIGH Q CAVITY;

ESTIN, A. J.

CPEM DIGEST 3p JUNE 1972.

DIRECT INTEGRATION OF THE FIELD EQUATIONS FOR ELECTROACOUSTIC

TRANSDUCERS;

YAGHJIAN, ARTHUR D. IEEE LETTERS, V62 N6 P858-859 JUNE 1974.

ANALYSIS OF A SIX-PORT JUNCTION FOR MEASURING, V, I, A, B, Z, GAMMA, AND

PHASE;

HOER, C. A.; ENGEN, G. F. PROCEEDINGS OF CONFERENCE OF THE INTERNATIONAL MEASUREMENT

CONFEDERATION (6TH), DRESDEN (GERMANY), 17-23 JUNE 1973, ACTA IMEKO, VI P213-

222 1973.

AUTOMATED CALIBRATION OF DIRECTIONAL -COUPLER-BOLOMETER-MOUNT

ASSEMBLIES;

ENGEN, GLEN F. PROCEEDINGS OF IEEE-MTT-S MICROWAVE SYMP. PALO ALTO, CALIF.

12-14 MAY 75, P95-97 MAY 75.

THE WAVE-HOP FIELDS FOR AN INCLINED DIPOLE OVER A SPHERICAL EARTH

WITH AN ANISOTROPIC IONOSPHERE;

LEWIS, RICHARD L.

48p OT-ITR-RR5, OCTOBER 1970.

EVALUATION OF REFLECTIVITY LEVEL OF ANECHOIC CHAMBERS USING

ISOTROPIC, 3-DIMENSIONAL PROBING;

CRAWFORD M. L.

PROCEEDINGS OF INTERNATIONAL IEEE/AP-S SYMPOSIUM, ATLANTA,

GA. 10-12 JUNE 1974, DIGEST, P28-34.

SCATTERING-MATRIX DESCRIPTION AND NEAR-FIELD MEASUREMENTS OF

ELECTROACOUSTIC TRANSDUCERS;

KERNS, D. M.

JNL. OF THE ACOUSTICAL SOCIETY OF AMERICA, V57 N2 P497-507

FEBRUARY, 1975.

APPLICABILITY OF SPEECH BANDWIDTH COMPRESSION TECHNIQUES IN MINE

ELECTROMAGNETIC COMMUNICATIONS;

MANNEY, C. H. NBS-10728, 25p 1972.

842
SYSTEM FOR MEASURING PEAK PULSE POWER USING SAMPLING AND COMPARISON

TECHNIQUES;

HUDSON, PAUL A.; ECKLUND, WARNER L.; ONDREJKA, ARTHUR R. PAT-APPL-233 002, PATENT-3 239 758, 2P 1962.

IMPROVED GEARING FOR ROTARY-VANE ATTENUATORS;

FOOTE, WILLIAM J.; HUNTER, RONALD D.

REVIEW OF SCIENTIFIC INSTRUMENTS, V43 N7 P1042-1043 JULY 72

A NEW FLANGE DESIGN FOR O-RING SEALS;

FOOTE, WILLIAM J.

REV. SCI INSTRU. lp AUGUST 1970.

NON-RECIPROCAL DEVICES USING SOLID STATE MAGNETOPLASMAS AT MILLIMETER AND

SUBMILLIMETER WAVELENGTHS;

KANDA, M.

NASA-CR-122923, 169p 1971.

IMPROVEMENTS TO THE NBS RF PEAK-PULSE POWER STANDARD;

SIMPSON, PHILIP A.; ONDREJKA, ARTHUR R. PRESENTED AT THE INSTRUMENT SOCIETY OF AMERICA SILVER JUBILEE

INTERNATIONAL CONFERENCE AND EXHIBIT, 26-29 OCTOBER 70, PHILADELPHIA, PA.

PAPER 709-70, PL-7.

THE APPLICATION OF SEMICONDUCTORS TO QUASI- OPTICAL ISOLATORS FOR USE AT

SUBMILLIMETER WAVELENGTHS;

HAYES, R. E.; KANDA, M. ; MASCLET, G.; MAY, W. G.

NASA-CR-86379, 61p 1970.

THE APPLICATION OF SEMICONDUCTORS TO QUASI- OPTICAL AND WAVEGUIDE

ISOLATORS FOR USE AT MILLIMETER WAVELENGTHS;

HAYES, R. E.; KANDA, M. ; MASCLET, G. ; MAY, W. G. NASA-CR-105980, 68p 1969.

EMERGENCY LOCATOR TRANSMITTERS. EFFECTIVE RADIATED POWER LEVELS AND TECHNIQUES

OF DETERMINING EFFECTIVE RADIATED POWER;

TAGGART, H. E.; WORKMAN, J. L.; NELSON, R. E. NBS-9769, 59p 1971.

BATTERIES USED WITH LAW ENFORCEMENT COMMUNICATIONS EQUIPMENT:

CHARGERS AND CHARGING TECHNIQUES;

SCOTT, W. W. JR.

NBS-10732, 49p 1973.

RADIO FREQUENCY COAXIAL AMMETER WITH THERMAL COMPENSATION;

SCOTT, WINSTON W. JR.

PAT-APPL-4 372, PATENT-3 609 541, 7P 1970.

NEW COAXIAL RF-DC AMMETER;

SCOTT, WINSTON W. JR.

IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, IM-19

N4 P318-323 NOVEMBER 1970.

843
DEJECTION OF HUMAN INTRUDERS BY LOW FREQUENCY SONIC INTERFEROMETRIC

TECHNIQUES;

STOLTENBERG, ROBERT E. NBSIR-74-364, 99p 1974.

RF NULL DETECTOR NBS/SND;

STOLTENBERG, ROBERT E. NBSIR-73-302, 89p 1973.

BATTERIES USED WITH LAW ENFORCEMENT COMMUNICATIONS EQUIPMENT;

JESCH, R. L. BERRY, I. S.

NBS-10722, 43p 1971.

A PRECISION, HIGH FREQUENCY CALIBRATION FACILITY FOR COAXIAL

CAPACITANCE STANDARDS;

JONES, R. N. ; HUNTLEY, L. E. NBS-TN 386 33p MARCH 1970.

THE 1973 INTERNATIONAL MICROWAVE SYMPOSIUM;

WAIT, DAVID F. ; BEATTY, ROBERT W.

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, MTT21

N12 P747-751 DECEMBER 1973.

SPECTRAL DENSITY ANALYSIS: FREQUENCY DOMAIN SPECIFICATION AND

MEASUREMENT OF SIGNAL STABILITY;

HALFORD, DONALD; SHOAF, JOHN H.; RISLEY, A. S.

ANNUAL SYMPOSIUM ON FREQUENCY CONTROL (27TH), CHERRY HILL,

N.J. 12-14 JUNE 1973, P421-431 (ELECTRONIC INDUSTRIES ASSOCIATION,

WASHINGTON, D.C. 20006).

MICROWAVE SPECTRAL TABLES. VOLUME III. POLYATOMIC MOLECULES WITH

INTERNAL ROTATION;

WACKER, PAUL F.; CORD, MARIAN S.; BURKHARD, DONALD G.; PETERSEN, JEAN D. KUKOL, RAYMOND F. NBS-MN-70-V0L-3. 3, 276p 1969.

INFRARED-MICROWAVE FREQUENCY SYNTHESIS DESIGN: SOME RELEVANT

CONCEPTUAL NOISE ASPECTS;

HALFORD, DONALD.

PROCEEDINGS OF FREQUENCY STANDARDS AND METROLOGY SEMINAR,

QUEBEC, CANADA, 30 AUGUST - SEPTEMBER 1971, P431-466 1972.

THEORY OF ADJOINT RECIPROCITY FOR ELECTROACOUSTIC TRANDUCERS;

YAGHJIAN, ARTHUR D. NBSIR-73-329, 79p 1974.

APPLICATION OF A NON-IDEAL SLIDING SHORT TO TWO-PORT LOSS

MEASUREMENT;

WEIDMAN, M. P.; ENGEN, G. F. NBS-TN-644, 33p 1973.

844
HIGH QUALITY QUARTZ CRYSTAL OSCILLATORS: FREQUENCY DOMAIN AND TIME DOMAIN

STABILITY;

BRANDENBERGER, HELMUT; HADORN, FREDERIC; HALFORD, DONALD; SHOAF,

JOHN H. PROCEEDINGS FREQUENCY CONTROL ANNUAL SYMPOSIUM (25TH)

ATLANTIC CITY, N.J. 26-28 APRIL 1971 P226-234.

MODULATION FACTOR STANDARD

ARTHUR, M. G.

NBS-TN-1016, SEPTEMBER 1979.

ELECTROMAGNETIC BOUNDARY -VALUE PROBLEMS BASED UPON A MODIFICATION OR RESIDUE

CALCULUS AND FUNCTION THEORETIC TECHNIQUES;

MONTGOMERY, JAMES P; CHANG, DAVID C.

NBS-MN-164, AUGUST 1979

WAVEFORM SAMPLER;

LAWTON, ROBERT A.; ANDREWS, JAMES R. PAT-APPL-670-468, PATENT-4 030 840, 19P 1976.

THE MEASUREMENT OF LUMPED PARAMETER IMPEDANCE;

JONES, R. N.

NBS/NM -141, 1974

845
ELECTROMAGNETIC NOISE METROLOGY

THE MEASUREMENT OF NOISE PERFORMANCE FACTORS: A METROLOGY GUIDE;

ARTHUR, M. G.

NBS/MN-142, 203p 1974.

A PRECISION HF-NOISE POWER MEASUREMENT SYSTEM;

ARTHUR, M. G.

INSTR SOC. AMER. ISA-70 CONF 5p OCTOBER 1971.

WR15 THERMAL NOISE STANDARD;

DAYWITT, W. C; FOOTE, W. J.; CAMPBELL, E. NBS-TN-615, 158p 1972.

AN IMPROVED SOLID STATE NOISE SOURCE;

KANDA, MOTOHISA.

PROCEEDINGS OF THE INSTITUTE OF ELECTRICAL AND ELECTRONICS

ENGINEERS-MTT-S MICROWAVE SYMP. CHERRY HILL, N.J. JUNE 14-16, 1976, P224-226

JUNE 1976.

STABILITY OF SILICON AVALANCHE NOISE DIODES AT K AND Ka BANDS;

KANDA, MOTOHISA; LARSON, T. R. CPEM DIGEST, 5p JUNE 1976.

A MEASURE FOR THE STABILITY OF SOLID STATE NOISE SOURCES;

KANDA, MOTOHISA.

PROCEEDINGS OF IEEE MICROWAVE THEORY AND TECHNIQUE SYMP. INT. (1975),

PALO ALTO, CALIF. 12-14 MAY 75, P315-317 1975.

STUDY OF ERRORS IN ABSOLUTE FLUX DENSITY MEASUREMENTS OF CASSIOPEIA A;

KANDA, MOTOHISA.

NBSIR-75-822, 35p 1975 AND IEEE TRANS. I & M SEPTEMBER 1976.

AN AUTOMATED PRECISION NOISE FIGURE MEASUREMENT SYSTEM AT 60 GHZ;

BOYLE, D. R.; CLAGUE, F. R.; REEVE, G. R. ; WAIT, D. F. ; KANDA, M.

IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VIM21

N4 P543-549 NOVEMBER 1972.

EVALUATION OF SIGNAL PLUS NOISE DETECTION ERROR IN AN ENVELOPE

DETECTOR WITH LOGARITHMIC COMPRESSION;

ESTIN, A. J.; DAYWITT, W. C.

PROCEEDINGS OF CONFERENCE ON PRECISION ELECTROMAGNETIC

MEASUREMENTS, OTTAWA, CANADA, 26-29 JUNE 1978 (IEEE CAT. NO. 78CH 1320-1 IM,

CPEM DIGEST P103-104 1978).

G/T MEASUREMENT ERRORS WITH RADIO STARS;

DAYWITT, W. C; KANDA, M.

PROCEEDINGS OF IEEE INT. ANTENNAS AND PROPAGATION SYMP.

URBANA/CHAMPAIGN, ILL. 2-4 JUNE 1975, SESSION 20, P460-463 1975.

CONSIDERATIONS FOR THE PRECISE MEASUREMENT OF AMPLIFIER NOISE;

WAIT, DAVID F.

NBS-TN-640, 133p 1973.

846
COMPARING FREQUENCIES

;

HALFORD, DONALD.

PHYSICS TODAY, V26 N2 P15 FEBRUARY 1973.

A STUDY OF THE MEASUREMENT OF G/T USING CASSIOPEIA A;

WAIT, D. F.; KANDA, M. ; DAYWITT, W. C; MILLER, C. K. S.

NBSIR 74-382, 202p OCTOBER 1974.

A REFERENCE NOISE STANDARD FOR MILLIMETER WAVES;

DAYWITT, W. C.

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VMTT21

N12 P845-847 DECEMBER 1973.

847
SUBJECT INDEX

Absolute ampere

Absolute electrical units

Absolute farad

Absolute ohm

Absolute resistance

Absolute volt

AC-DC converter

AC-DC difference

AC-DC transfer instruments

AC-DC transfer standards

Admittance measurement

Admittance standards

Alloys for resistors

Ampere

As-maintained electrical units

Automated bridge

Automated calibration; ac-dc difference

Automated comparator; ac-dc difference

Automated potentiometer

Automated resistance comparator

Automated resistance measurements

Automated standard cell comparator

Automated voltage measuring system

Automatic bridge

Automatic comparator for TVCs

Automatic intercomparison, voltage

Avogadro constant

Bridge

Calculable capacitor

Calibration, low frequency

Calibration, ac voltage

Calibration, ac-dc difference

Calibration, thermal current converters

Calibration, thermal voltage converters

Calibration, capacitance

Capacitance

Capacitors with dielectric films

Capacitors, calculable

Capacitors, gas dielectric

Capacitors, mechanical tests

Capacitors, temperature compensated

Codata

Comparator

Comparison, inductance and capacitance

Compensated capacitors

Control charts for standard cells

Converter, ac-dc

17,117,124,128,149

16

144

16,144,149

367

16

692

640,667,672,682,542,537

598

636

463

475,521

380

3,15,117,124,128,149

15

162

542,640

537

339

339

346

241,251

237

461

631

332,333

17

463,475,479,491,533,677

144,149,260,367,471

107

548,646

542,640

669

631

399

151,399,491,509

525

525

412,455

412

504

18

537,598,631,661,667,682

533

504

293

692

848
Subject Index -- continued

Cross capacitor

Current balance

Current comparator

Current transfer

Current, AC

Designs for standard cell groups

Digital voltmeter

Direct current comparator

Direct reading ratio set (DDR's)

Elecrical units

Electrical measurement research

Electrical units

Electrometers

Farad

Fine structure constant

Fundamental constants

High DC voltage

Impedance

Inductance

Inductive voltage dividers, calibration

Interlaboratory transfer, voltage

International comparison of volt

Josephson devices

Josephson effect

Josephson volt

Laboratory electrical units

Least squares adjustment

Legal electrical units

Manganin

Measurement assurance programs

Measurement assurance, capacitance

Mult ij unction thermal converter (MJTC)

National measurement system, electricity

NBS calibration, AC voltage

NBS services

NBS standards, electrical

NBS unit of resistance

Ohm

Pellat ballance

Peltier effect

Potentiometer

Programmable automatic bridge

Proton gyromagnetic ratio

Proton mass

Proton, gyromagnetic ratio

144

128,149

677

260

661,667

303

646,656

334

475,483

32

21

3

153

17,124,144

17,170,181,233

3,11,15

247

150,521

150

499

322

227

231,232

208,211,215,233

260

15

15,18,21

33

382

312

399

646,662,672

23

548

69,107

107

215,221,227,256,367

3, 15,144,149

124,128

393

232,334

461

6,260

17

17

849
Subject Index -- continued

Proton, magnetic moment

Quadrature bridge

Quantized hall resistance

Quantum hall devices

Quantum hall effect

Regional volt maintenance

Resistance

Resistive divider switching

Resistive ratio standard

Resistor calibration

Resistors

RMS voltage calibration

RMS voltmeter

Rydberg constant

Sampling voltmeter

SI ampere

SI electrical units

SI units

Solid state voltage reference

Squid

Standard capacitance

Standard capacitors

Standard cell calibration

Standard cell comparator

Standard cell construction

Standard cell control charts

Standard cell EMF

Standard cell enclosure

Standard cell intercomparison

Standard cell statistics

Standard cell temperature coefficient

Standard cell temperature coeficient

Standard cells

standard cells

Standard cells

Standard resistance

Standard resistor calibration

Standard volt

Standards, electrical

Standards, AC voltage

Standards, AC-DC difference

Standards, admittance

Standards, impedance

Surface films in gas capacitors

Temperature coefficient of resistance

Thermal current converters (TCC)

Thermal voltage converter (TVC)

17

479

3

162

159,165,170,174,185,197

327

3,159,162,165,170,185,197,

339,350,371,393,677

344

247

346

380

646

656

17

656

117,124,128

3,15

15

237

211,677

399

412,525

260,293

241,251

335

293

280

256

332,333

293

272,264

280

272

293

322,332

159,165,170,181,197

339,350

208,215,233

149

548

636

521

521

455

382

667

631,640,646,652,661,672,682

850
Subject Index -- continued

Thermodynamics of standard cells 272

Thermoelements (TE's) 640

Thermoelements, AC-DC difference 599

Thompson and lampard capacitor 144

Traceability of electrical energy 39

Traceability of voltage 38

Transfer standard, capacitance 399

Transportable capacitance standard 399

Transportable standard cell enclosure 334

TVC intercomparison 636

Varactor null detector 491

Volt 3,15,208,211,215,218,231,293,

327,332,333,334,355,548,656,661

Von Klitzing effect 174

Zener diodes 322

Zener voltage reference 237

851
AUTHOR INDEX

Auxier, L. M. TOT 32/

Baker, K. K.

inn 339

Belecki, N. B. n 771 23 , 321

Bower, V. h. i i 7 11/

Braudaway, D. W. 231 , 251 , 334

Brown, C. F.

nil 312

Lage, M. Ji. 17/, 1 CQ 1 £C 170 1QC 1Q7 1Z4, i_>y, ±oo , i/u, ibj, iy / Cameron, J. M. OAT 2y3

Cnilders, C. B. 1 £ 1 7 /

161 , 247

Crovini, L. 393 mi

Cutkosky, R. D. roc 1// i/r» 7 c £ 7 / / 7 £ 7 / /; 1 / 7c / ni crw c 7 i 525, 144, 149, 256, 344 , 36 7 , 46 1,475, 491, 504, 521

Denenstein, A. 7 7 1 777 777 231 , 232 , 233

Dunree, B. L. 23 , 355

Dunn , A . r

Tin nn-i 332 , 333

Dziuba, R. F. 1CH 71 1 7/7 77fl nrp £"7*7 1 £C 159, 211, 24 7, 339, 355 , 67 7 , 165

hi i eke , W. G. , Jr. 7/1 7 n 7 11 7 777 777 241, 293, 312, 32/, 22/

IT A o 1 A U IT r lela , B . r

1 CQ 1 A 7 1 fit 7 fi Q 711 7 1 R 7 77 TCt 7£ft 7ft/, ft ^ ft i jy , loz , i o_> , zuo , z 1 1 , z i j , z J / ,Zjo,_ou,zd4,ojd

r innegan, 1 . r . 711 71 C 771 777 777 211, 215, 231, 232, 233

rlacn, D. R. C / 7 £77 £07 542 ,11 , 692

Free, G. 7 n 7 39 / oalaknova, u. r. ft7ft 636

H « mm A T7 T? bard , li . r . 7 en 3 jU

dirvin, b. M. 1 Trt 1 OC 1 /0 , 185

Hamer , W. J. 7 A /. 7 7 7 Z04 , Z /

Harkness, S. 0 JO

nermacn, r . L

1f"\7 £7£ £77 ££.1 10/ , 636 , 6/2 , 661

Hesterman, V. W. 7no ZUo

Hirayama, H. 7 7 7 £ 7 & 333 , 636

Jones, b. K* , Jr

ft 117

T _____ _ T% VT Jones, R. N. c n 7 5U/

Kmard, J. R. C77 537

Kirby, C. G. 7 n 7 7 C\ 7 393 , 393

Kleiman, R. E. O / 1 1 C 1 241 , 251

Kusters, N. L. O O / 334

Lampard, D. G. 525

Langenberg, D. N. 233

Laubitz, M. J. 393

Leedy, T. F. 312

Lentner, K. J. 542,640

Levy, C. R. 412

Lisle, R. V. 499

MacMartin, M. P. 334

Martin, P. 636

Marzetta, L. A. 692

Moore, B. R. 312

852
Author Index -- continued

Morrow, J. J. 397,346

Muciek, A. 533

Murayam, Y. 333

Olsen, P. T. 6,117,124,128,260

Peterson, C. 379

Petersons, 0. 23

Phillips, W. D. 6,117,124,128

Rozdestvenskaya, T. H 636

Schoenwetter , H. K. 548,646

Shaw, R. M. 332

Shields, J. Q. 455,463,471

Skapars, A. 264

Sloggett, G. J. 332

So, E. 455

Song, J. 6

Sullivan, D. B. 677

Taylor, B. N. 3, 11, 15,227

Toots, J. 215

Tremaine, S. G. 161

Vincent, G. D. 335

Wells, T. E. 350

Wilbur-Ham, J. 333

Williams, E. R. 6,117,124,128,260,

Williams, E. S. 537,598,631,636,667,682

Zapf, T. L. 499

853
NBS-114A (REV. 2-8C) U.S. DEPT. OF COMM. 1. PUBLICATION OR 2. Performing Organ. Report No. 3. Publication Date BIBLIOGRAPHIC DATA

REPORT NO.

October 1985 SHtET(See instructions) NBS/SP-705

4. TITLE AND SUBTITLE

Precision Measurement and Calibration: Electricity

Selected NBS Publications on The Realization and Maintenance of the Fundamental

Electrical Units and Related Topics

5. AUTHOR(S)

A. 0. McCoubrey -Editor

6. PERFORMING ORGANIZATION (If joint or other than NBS. see instructions)

NATIONAL BUREAU OF STANDARDS

U.S. DEPARTMENT OF COMMERCE

GAITHERSBURG, MD 20899

7. Contract/Grant No.

8. Type of Report & Period Covered

Final

9. SPONSORING ORGANIZATION NAME AND COMPLETE ADDRESS (Street, City. State. ZIP)

Same as item 6.

10. SUPPLEMENTARY NOTES

Library of Congress Catalog Card Number: 85-509580

J Document describes a computer program; SF-185, FlPS Software Summary, is attached.

11. ABSTRACT f A 200-word or less factual summary of most significant information. If document includes a significant

bibliography or literature survey, mention it here) Selected publications of the National Bureau of Standards technical staff in the

field of electricity were first compiled in 1962 as a volume of the NBS Precision

Measurement and Calibration Series (Electricity and Electronics, Handbook 77,

Volume I); this compilation was extended in 1968 by the compilation of an additional volume in the Precision Measurement and Calibration Series

(Electricity-Low Frequency, NBS Special Publication 300, Volume 3). The present

volume, a further extension of these earlier compilations of selected publications

in the field of electricity, includes 66 more recent papers by NBS authors and 16

abstracts of closely related papers by authors from other organizations. In view

of the expansion of measurement technologies used in electricity and electromagnetism it has been necessary to reduce the range of topics for the selection of

papers in the new compilation. In this connection an emphasis has been placed

upon the realization and maintenance of fundamental electrical units and the

related scientific advances, particularly in quantum physics. However, in the

interest of completeness, three appendices also provide up-to-date bibliographies

of publications by NBS authors in different areas of electromagnetism.

(This book is a sequel to NBS Handbook 77-Vol. 1(1961) and NBS SP 300-Vol. 3

(1968).

12. KEY WORDS (Six to twelve entries; alphabetical order; capitalize only proper names; and separate key words by semicolons)

Capacitance standards; electrical calibrations; electrical measurements; electrical

standards; electrical units; Josephson standards; quantum standards; resistance

standards; standard cells; voltage standards.

13. AVAILABILITY

[X] Unlimited *

| | For Official Distribution. Do Not Release to NTIS

[x] Order From Superintendent of Documents, U.S. Government Printing Office, Washington, D.C. 20402.

~Jt Order From National Technical Information Service (NTIS), Springfield, VA. 22161

14. NO. OF

PRINTED PAGES

858

15. Price

USCOMM-DC 6043-P80 * U.S. GOVERNMENT PRINTING OFFICE 1985 461-135/36653
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