NATL INST OF STAND S. TECH

AlllDb T7 CI134

NBSIR 83-1689

MBS

PUBLICATIONS

APPROXIMATE FORMULAS

FOR THE FAR FIELDS AND GAIN OF

OPEN-ENDED RECTANGULAR

WAVEGUIDE

Arthur D. Yaghjian

National Bureau of Standards

U.S. Department of Commerce

Boulder, Colorado 80303

May 1983

-QC

100

,U56

01-1609

1903

C . 2

NBSIR 83-1689

APPROXIMATE FORMULAS

FOR THE FAR FIELDS AND GAIN OF

OPEN-ENDED RECTANGULAR

WAVEGUIDE

Arthur D. Yaghjian

Electromagnetic Fields Division

National Engineering Laboratory

National Bureau of Standards

U.S. Department of Commerce

Boulder, Colorado 80303

May 1983

U.S. DEPARTMENT OF COMMERCE, Malcolm Baldrige. Secretary

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director

CONTENTS

Page

I. INTRODUCTION 1

II. DERIVATION OF FAR-FIELD FORMULAS 3 A. The E-Plane Pattern 4

B. The H-Plane Pattern 6

1) The E-Field Integration Method for the H-Plane Fields... 6

2) The Fringe Current Method for the H-Plane Fields 8

III. DETERMINATION OF ON-AXIS GAIN 14

IV. CONCLUSION 15

APPENDIX I 17

APPENDIX II 20

ACKNOWLEDGMENT 23

REFERENCES 24

i i i

Approximate Formulas for the Far Fields and Gain of

Open-Ended Rectangular Waveguide

Arthur D. Yaghjian

Electromagnetic Fields Division

National Bureau of Standards

Boulder, Colorado 80303

Approximate formulas are derived for the far field and gain

of standard, open-ended, unflanged, rectangular waveguide probes

operating within their recommended usable bandwidth of frequencies. (Such probes are commonly used in making probe-corrected

near-field antenna measurements.) The formulas, which yield

forward far-field power patterns and on-axis gains of X-band and

larger waveguide probes to within about 2 dB and 0.2 dB accuracy,

respectively , assume ( s i n <> - cos^) azimuthal angular dependence

and an E-plane pattern given by the traditional aperture integration of the TE 10 mode E- and H-fields in the Stratton-Chu

equations. The H-plane pattern is estimated by two different

methods. The first, and less accurate, method uses a purely Efield aperture integration to estimate the H-plane pattern. The

second, more accurate, method uses an electric field integral

equation (EFIE) to show that fringe currents near the shorter

edges of the guide can be wel 1 -approximated by isotropic line sources. The amplitude of these line sources is then determined

more simply by equating the total power radiated into free space

to the net input power to the waveguide.

Key words: edge diffraction, fringe current, physical optics,

radiation pattern, rectangular waveguide.

I. INTRODUCTION

Most near-field antenna measurements are made using open-ended, unflanged,

rectangular waveguide probes. They are simple, rugged, and inexpensive to

produce (or reproduce), they have a broad far-field power pattern that

1
contains no nulls in the forward hemisphere over their recommended usable

bandwidth, they scatter relatively little radiation back to the test antenna

being measured, and they have a respectable gain of about 6-8 dB. However,

unlike far-field measurements where only the on-axis gain of the probe is

required, near-field measurements require, in general, the complex far field

in the forward hemisphere of the probe in order to compensate for its presence

in the near field. Specifically, for each microwave frequency at which the

test antenna is to be measured, the appropriate standard rectangular waveguide

probe is chosen and its far-field pattern, gain, and reflection coefficient

are measured as a necessary part of the complete near-field measurement

procedure.

The reflection coefficient can be measured accurately and rapidly over

the waveguide bandwidth using, e.g., an automatic network analyzer, but the

measurement of probe pattern and gain is a rather tedious, time-consuming

operation, especially when the fields of the test antenna are desired at

several frequencies for which the probe has not been previously calibrated.

To alleviate this chore of measuring the probe far fields, at least when

extremely high accuracy is not required, we have derived reasonably accurate

formulas for the far-field pattern and gain of open-ended, standard rectangular waveguides over their recommended usable bandwidths. For example, comparison with measured data indicates that equations (2) and (8) inserted into (1)

approximate the forward far-field power pattern and on-axis gain of X-band

(WR-90) and larger standard waveguide to within about 2 dB and 0.2 dB accuracy, respectively (see figures 3, 6, and 7), the maximum pattern error occurring near 90° off boresight. This accuracy represents a significant improvement over previous approximate formulas derived either entirely by aperture

integration with the Stratton-Chu formulas [1, Sec. 8.14; 2], or by a first

order geometrical edge diffraction technique [3]. Both of the previous

methods yield far-field power patterns several dB too high near much of the H- plane cut, and predict on-axis gain values which are typically 2 to 3 dB too

low. For Ku-band (WR-62) and smaller standard waveguides, the approximate

formulas derived in this paper may become less accurate because the wall

thickness of these smaller waveguides becomes an appreciable fraction of their

hei ght

.

2
II. DERIVATION OF FAR-FIELD FORMULAS

We take a semi-empirical approach to approximate the far fields and absolute gain of open-ended waveguide. In principle, one could obtain a numerical

solution for the fields everywhere by solving an electric field integral equation (EFIE) for the currents over the surface of the semi -infinite waveguide. And, indeed, we use this EFIE approach (formulated in Appendix I) in

Sec. II-B-2 to estimate the fringe current contribution to the H-plane

fields. However, we do not want to finally rely on numerical integral equation solutions because for 3-D problems like the open-ended rectangular

waveguide, they demand an exorbitant amount of computer time and storage to

achieve the desired accuracy, they are not nearly as simple to use as the

approximate formulas, and they inspire less physical insight.

The geometry of the open-ended rectangular waveguide is shown in figure

1. The inner dimensions of the waveguide are given by width a and height b.

The perfectly conducting waveguide walls are assumed to have a thickness small

enough compared to the smaller dimension b so that the finite thickness of the

walls can be ignored. The origin 0 of the xyz rectangular coordinate system

is chosen at the center of the open end (aperture or mouth) of the waveguide,

and the spherical coordinates of the position vector r to any point in space

is denoted by ( r , 0 , 4> ) . The waveguide operates at a frequency, f, that lies

within the recommended usable bandwidth of the TE^g mode (electric field in

the y direction). For example, for standard X-band waveguide, the recommended

usable bandwith ranges from 8.2 to 12.4 GHz, or equivalently the free-space

wavelength, ranges from 1.6a to 1.06a, a = 2.286 cm.

Consider the far field of the waveguide expressed as a sum of spherical

multipoles located at the origin 0. Because the transverse dimensions of the

waveguide are less than a wavelength, only the multi poles of lower order

azimuthal (<t>) dependence will contribute significantly to the far field.

Moreover, because all but the first-order ( s i n <$> - cos4>) dependent multi poles

have a null in the on-axis (z) direction, one would expect a significant con- tribution only from these first-order multi poles. Under this assumption and

the symmetry of the rectangular waveguide excited by the TE^g mode, the far

3
fields of the open-ended waveguide can be expressed approximately in the following simple form [1, Sec. 7.11]:

ikr

E(r-»°°) = - [E ( 0 ) si n4>e + E(0)cos<|>e ] (la)

kr E 0 H <t>

ikr

H(r>®) = — -=- [E_( 0) si n<|)e - E (0)cos<i>e ] , (lb)

kr L t (pH 0 o

where e' lu)t time dependence has been suppressed (w = 2%f, k = 2u/\), and Z

Q is

the impedance of free space. Equations (1) immediately reduce the problem of

finding the far fields to that of finding the E-plane and H-plane patterns,

E (0) and E (e), respectively. Indication that equations (1) represent the

far fields of the open-ended waveguide to a rather good approximation in the

forward hemisphere can be seen from figure 2 which plots the measured power

patterns of an X-band waveguide at 9.32 GHz for $

= 0, 45, and 90 degrees, and 12 2

compares the 45-degree pattern with j

[lE^I + I

E

H

l ], i.e., with the power

pattern predicted by (1) from the measured E- and H-plane patterns. In particular, the (^-dependence of (1) conforms more closely to the measured power

patterns than the "separable dependence" obtained from aperture integration

with the Stratton-Chu formulas [2] or through first-order geometrical theory

of edge diffraction [3]. The approximation (1) would not, however, be

expected to give very accurate polarization ratios for wide 0 angles near <j> = 45°.

A. Hie E-Plane Pattern

The E-plane pattern, E (0), is predicted quite accurately by inserting

the E- and H-fields of the propagating TE 10 mode into the Stratton-Chu formulas [l,Sec. 8.14] and integrating over the mouth of the open-ended waveguide

[2]. Risser explained in [2], and we prove in Appendix I, that this StrattonChu integration of the TE^q mode is equivalent to integrating over the

physical optics current, i.e., the truncated interior surface currents K 10 of

the propagating TE^ mode (incident plus reflected). In other words, the

resulting far fields obtained from the Stratton-Chu formulas neglect only the

4
evanescent mode currents inside the waveguide and the surface currents on the

outside of the waveguide. Because the sum of both these currents equal the

total surface current less the physical optics current, we can simply say that

the Stratton-Chu formulas applied to the TE-^g mode neglect only the Ufimtsev

"nonuniform" or "fringe" currents [12].

Although these fringe currents can contribute appreciably to the absolute

value of the field, they have a minor secondary effect in the broad E-plane

pattern which remains well above -10 dB for all angles 9. Thus, for E^(0) we simply use the classical result of the aperture integration with the StrattonChu formulas [2]

E

e (9)

[1 + £

cose + r ( 1 - cose)] sin(-^- sine)

Ae

[l+|+r(l-£)] T sine

( 2 )

The normalized propagation constant p/k for the TE 10 mode equals A - and the reflection coefficient of the TE^g mode from the end of the waveguide

is denoted by r. The constant A which remains arbitrary at this point in the

derivation is eventually related to the amplitude of the incident TE 10 mode

through equation (9). The factor [1 + + r (1 -

-j^-)] in the denominator of

(2) is included merely to simplify the normalization at e = 0.

Since (2) could be obtained by integrating the truncated TE 10 surface

currents, it remains a valid approximation for e in the back as well as the

forward hemisphere. However, (2) becomes a better approximation in the

forward hemisphere when r is set equal to zero because zeroing r tends to

compensate in the forward-hemisphere E-plane for the neglected fringe cur- rents. Thus, whenever (2) is used in the region of most interest, i.e., the

forward hemisphere, r will be assumed zero. If (2) is used in the back

hemisphere, a more accurate pattern results by including the reflection

coefficient beyond e equal to about 105°.

In figure 3 the amplitude and phase of E ( 0) in the forward hemisphere is

plotted from (2) and compared with NBS (National Bureau of Standards) measured

data for X-band waveguide at 9.32 GHz and L-band (WR-650) waveguide at 1.0

5
GHz. The close agreement between the theoretical and experimental curves was somewhat unexpected for the L-band guide because the 1 GHz frequeny is

approaching the 0.908 GHz cut-off frequency and lies well below the lowest

recommended usable frequency of 1.12 GHz.

B. The H-Plane Pattern

The H-plane pattern, E (e), is much narrower than the E-plane pattern

and, consequently, it is more strongly influenced by the fringe currents

neglected in the Stratton-Chu aperture integration. Specifically, the

Stratton-Chu integration with the TE^g mode yields an H-plane pattern much

broader than the measured pattern. And this is true whether the reflection

coefficient r of the TE^g mode is included or not. Thus we are left with

finding a method that will predict accurately and simply the H-plane pattern

of open-end waveguide over the recommended usable bandwidth.

Two separate methods will be developed for determining the H-plane pattern. The first method, referred to as the E-field integration method, does

not require the reflection coefficient r, but produces a valid H-plane pattern

over the forward hemisphere only, and erroneously becomes exactly zero at 9

=

90°. The second, a more complicated method referred to herein as the fringe

current method, does require the reflection coefficient and yields an accurate

H-plane pattern for all e--front and back hemisphere. This second method also

yields slightly more reliable values of on-axis gain.

1) The E-Field Integration Method for the H-Plane Fields

As explained above, aperture integration of the Stratton-Chu formulas

with the electric and magnetic fields of the TE 10 mode neglects the fringe

currents and thereby produces much too broad an H-plane pattern. However, the

Stratton-Chu formulas are not the only expressions that can be used to predict

the far fields from an aperture integration of the fields of the TE 10 mode.

In particular, the electric or magnetic far field in the forward hemisphere

can be expressed respectively as a double Fourier transform of the electric or magnetic field alone over the infinite plane just in front of the mouth of the

6
waveguide. Concentrating on the electric field, for reasons that will become

clear in the next paragraph, we express the H-plane far field as [4]

E( r-*» ,0 ,<j>=0)

0<9<Tt/2

ikr ,2

e k_

kr 2 7i e

r

x e

z

CD 00 X I I

-ikx'sine

E(R')e dx'dy' , (3)

where R

1 = x'e

x + y'e^ is the integration vector from the origin 0 to an area element dx'dy' in the plane of integration.

Now the electric and magnetic field (E 10 and H10 ) of the TE 10 mode 1S

given by

E 10

= E0

(1 + r)cos (T)e

y

(4a)

E H

10

=

• T

[(1 - r)

k

cos(-7)e

x + Tk?

(1+r)sin ^)e

z

;l • (4b)

0

where EQ

is the arbitrary amplitude of the incident TE^g mode. Note that the

E-field has but one component that goes to zero at the edges x = +a/2. This

suggests there will be little difference between the average x-variation of

the electric field just in front of the mouth of the waveguide and that of the

TE^g mode. In other words, for the purpose of determining a reasonably

accurate H-plane pattern, the E-field on the integration plane in (3) can be

approximated by the E^g of (4a) over the mouth or aperture of the waveguide

and zero outside. Performing this simple integration of (3) gives an approximate H-plane pattern valid in the forward hemisphere

E

H

( e ) = A

E (|)

2 cose o<e<!

k 3

cos(—^-si ne

)

t(f) 2

- Cosine) 2

]

(5)

The constant A in (5) is the same as in (2) because E (0) must equal E ( e

)

along the common

E axis 0=0. The same integration with EEH

10 does not and would

not be expected to yield an accurate E-plane pattern because the E-plane

7
pattern is determined by the y-variation of the aperture field which differs

significantly from that of the incident E^q mode field through diffraction

from the edges of the waveguide at y

= + b/2. Also, substituting H10 from

(4b) into the formula for the magnetic field corresponding to (3) yields a very poor approximation for both the E- and H-plane far fields. This, of

course, is no surprise because the H^q field does not become zero at any of

the waveguide edges.

Figure 4 compares the amplitude and phase curves for the H-plane fields

in the forward hemisphere calculated from (5), measured at NBS, and calculated

from the following Stratton- Chu expression for the H-plane pattern [2]

E

H (e) = A,

[(cose + f) + r(cose - £)] k - cos(—~si n9 j Hca

ne)

2

]

( 6 )

where Au = -ik 2 ab E /8.

The H-plane fields predicted by (5) agree well with the measured data even

near the L-band cut-off frequency, and they are much closer (on a linear

scale) to the measured values than those predicted by the result (6) of the

Stratton-Chu formulas.

2) The Fringe Current Method for the H-Plane Fields

Although the integrated E-field expression (5) for the H-plane pattern is

much more accurate in the forward hemisphere than the previous expressions

[2,3], it has a couple of shortcomings that the E-plane expression (2) does

not have. First, the expression (5) does not remain valid (or accurate) in

the back hemisphere because it was derived from (3) which applies only to the

front, free-space hemisphere. Thus, when the constant A

E is related to total

power radiated, the H-plane pattern in the back hemisphere must be estimated

and this estimation introduces an additional, albeit small, inaccuracy into

the determination of gain (see Sec. III). Secondly, the expression (5)

8
approaches zero like cos0 as 0 approaches 90°, and predicts too low an H-plane

pattern for 0 between about 70° and 90°. Fortunately for all near-field

antenna measurements, except those made with the probe extremely close to the

test antenna, the pattern of the probe beyond 70° does not intercept the test

antenna and thus does not affect the probe correction [5,6,7]. However, for

the sake of those rare exceptions as well as for slightly more reliable estimates of gain, we derive an alternative expression to (5) for the H-plane pattern that remains a good approximation for all values of 0.

To derive this alternative expression, we return to a formulation of the

open-ended waveguide problem in terms of surface currents. As mentioned

above, discussed by Risser in [2], and proven in Appendix I, the far fields of

the open-ended waveguide obtained by an aperture integration of the TE^q mode

fields in the Stratton-Chu formulas neglect only the fringe currents (total

current minus physical optics current, or equivalently, evanescent mode cur- rents plus the exterior surface currents). From the symmetry of the rectangular waveguide and TEjq mode, only y-directed currents contribute to the H- plane fields. Thus, equation (6) neglects only the fringe currents along the

x = +a/2 edges of the open end of the waveguide (provided the finite waveguide

wall thickness of the waveguide can be ignored). By estimating the contribution to the H-plane of these fringe currents and adding it to the Stratton-Chu

result (6), an alternative formula to (5) for the H-plane pattern is obtained

that remains a reasonably accurate approximation for all values of 0.

We first tried to obtain a good estimate of the y-directed fringe currents by looking at the x = +a/2 or -a/2 edge as part of the infinite edge of

a half plane. The exciting field was taken as the incident and reflected

plane wave produced by the incident and reflected TE 10 mode. Since (6) is an exact closed- form expression for the H-plane contribution from the physical

optics current (i.e., the truncated TE 10 mode current), the only additional

information we needed was the far fields of the fringe currents of the half

plane E-wave problem. For this problem, these fringe currents produce far

fields that are also expressible by exact, nonsingular, closed form expressions derived by Ufimtsev for the general wedge [12] and recast for the half

plane into simplified form in Appendix II. This latter result is worth emphasizing--the far-field diffraction pattern of the Ufimtsev fringe currents for

9
the E-wave incident upon the half plane is given exactly with no singularities* by the simple function (A8) involving only sines and cosines of the

observation angle and the angle of plane-wave incidence.

Appendix II shows that this half-plane fringe current correction works

very well for the open-ended parallel -pi ate waveguide excited by the TE^q

mode, but unfortunately for the open-ended rectangular waveguide, it did not

bring the predicted H-plane pattern to within the desired 1 or 2 dB accuracy

of the measured pattern at the wider angles of observations. The reason for

the greater inaccuracy in the case of the rectangular waveguide is simply that

the top and bottom edges of the rectangular waveguide strongly influence the

edge diffraction from the x = ±a/2 side edges, and produce fringe currents

significantly different from those of the half-plane problem. One may be able

to get a better estimate of the fringe current contribution by examining the

mutual interactions between the adjacent edges of the rectangular waveguide;

but such an analysis amounts to determining the diffraction coefficients for

the open corner problem, which eludes a closed-form solution.

We were finally able to obtain an accurate estimate of the fringe currents on the x = +a/2 sides of the rectangular waveguide from a numerical

solution to the EFIE applied to the open-ended rectangular waveguide. Although the detailed formulation of the EFIE solution is given in Appendix I,

just one major result of that solution is used here to modify the H-plane

pattern (6). Namely, the numerical EFIE solution revealed that the fringe

currents at x = ±a/2 contributed to the H-plane pattern like line sources that

radiate i sotropical ly with 0. That is, the contribution to the H-plane

pattern from the fringe current can be approximated simply by that of equal

isotropic line sources at (x = ±a/2, z =0), or specifically by

A

r

C

q

cos(^-|si ne J , (7)

*In making this statement, we are assuming that the angles of incidence and

scattering do not both equal the grazing (edge on) angle, 0 = a = + it in

(A8). For the plane waves composing the TE 10 mode in the waveguide, a is never too close to + n, when operating within recommended bandwidth.

10
where C

Q

is a positive real constant. Adding (7) to (6) then determines the

H-plane pattern derived by the fringe current method

(cose + £) + r(cose - £)

E H<e) = A, - (^sine)^

+ c. cos(-^si ne) ( 8 )

The constant A

H is defined under (6) and the H-plane constant A

E in (2)

is related to A by E (0) = E (0), or

H E H

a

e = V(|)

2

[(1 + r(! - !•)] + c0

}

. (9)

The positive real constant C

Q

, which depends on the frequency and the wave- guide dimensions, remains the one unknown in the fringe current expressions

(8) and (9). It can be determined in a number of ways, including directly

from the EFIE program discussed in Appendix II. A simpler way to find C

0

, however, is to equate the radiated power P

r determined from the far-field (1)

(with (2) and (8) inserted) to the total input power P Q

determined from the

TE^q mode fields (4), and solve the resulting quadradic equation for CQ

. From

(1) and (4), P Q

and P r are found to be, respectively.

a b |E I 2( 1- 1 r

1

2

) p/k

P o

= TTZ^ {10a)

P = — j

n

(IE ( 0)

1

2 + IE (9)

|

2

) si nede (10b)

r 2Z

0

k 0

Because the resistive loss in the waveguide walls is negligible, P r must equal

P . Upon equating (10a) and (10b), we find

11
a C (11a) ‘ q o 2b C

q o + Cq = °

or

b) - c a -3- a_a , nib)

"q

where a

q

, b

q

, and c

q are constants determined by integrals of the E- and H- plane patterns, (2) and (8). These integrals were evaluated numerically and

C

Q

from (lib) was determined for X-, C-, and L-band waveguides (WR-90, WR-187,

WR-650) . The on-axis gain G

0 is obtained directly from CQ through the

definition

c

~ b

3 * /(

G

=

o

47ir2|E(r^ } e=o)|

2

2 Z P o o

2-2

8*r |E|

2 B 2 IE lab £ (l-|r| ) o k

(12a)

from which we find, after inserting the on-axis far field from (2) or (8) into

(12a)

G

o

?

uk ab 2,2

8 |

(i-|rl)

7lb+t+r(l-f] (f) + C

o'

(12b)

Likewise, CQ can be written in terms of the on-axis gain by solving (12b) for

C

Q

in terms of G

Q

,

(12c)

where

c

e 5 r 1 + £

+ - f) i (|)

2

and

12
A

\

•jtk^ab

8 £

(l-lrl

2

)

*

For I r | (1-p/k) « (1+p/k), (12c) reduces to

(12d)

Since (12c) or (12d) are simple expressions for the constant C

Q as a function

of gain, and since the predicted gain G

Q from (12b) is plotted in figure 7 for

X- , C-, and L-band waveguides, C

Q

will not be plotted explicitly. The complex

reflection coefficients over the recommended usable bandwidth of the X-, C-,

and Ku-band waveguides, needed to determine C

Q

from (11), were obtained from

measurements with an automatic network analyzer at NBS. The amplitude and

phase of these measured r are shown in figure 5. (The L-band reflection

coefficient was estimated by scaling the given Ku-band data, since b/a for Land Ku-band waveguide are the same and L-band data was not immediately

available. Of course, some error is introduced by this scaling because the

wall thickness of Ku- and L-band waveguide do not scale like the inner

dimensions.)

Using these measured reflection coefficients and the C

Q

evaluated from

(11), the amplitude and phase of the H-plane pattern (8) is plotted in figure

6 for the forward hemisphere of X-band waveguide at 9.32 GHz and L-band wave- guide at 1.0 GHz. These predicted H-plane patterns (8) derived from the

fringe current method agree well with measured patterns even near e = 90° and.

though not shown in figure 6, with measured patterns over the back hemisphere

as well. The disadvantages of using (8) instead of the H-plane expression (5)

derived from the E-field method are the extra computations needed to evaluate

C

Q

and the need to supply the reflection coefficient r.

To determine C

Q

from (11), the far-field integrals in (10b) must be

evaluated over the full hemisphere. This was done numerically by simply

converting the integrals to summations and adding on a computer. The reflec13
tion coefficient in the E-plane (2) was assumed zero for e < 105° and taken as its measured value for e > 105° (see Sec. 1

1 -A )

.

III. DETERMINATION OF ON-AXIS GAIN

Two approximations have been derived for the far fields of open-ended

rectangular waveguides. The first uses equation (2) and (5), and the second

uses equations (2) and (8) for the E- and H-plane patterns in (1). This

section evaluates the absolute on-axis gain predicted by each approximation.

Because the resistive loss in the walls of the waveguide is negligible,

the total radiated power P r must equal the net input power P and the on-axis

gain (defined in (12a)) becomes equal to the on-axis directivity:

where P r is expressed by (10b) in terms of the E- and H-plane patterns. Using

(2) with r = 0 for the E-plane pattern and (5) for the H-plane pattern, (13) can be expressed explicitly as

cancels in the determination of gain from (14). If desired, the magnitude of

G

o (13)

for (2) and (5) can be written in terms of the coefficient E Q of the

incident TE^q mode by simply equating P Q

and P p

to get

( 15 )

where "Denom" denotes the denominator of the right side of (14).

14
It should be pointed out that the integral denominator in (14) was not

evaluated exactly as it is written to get the gain values labeled by eq (14)

in figure 7. As explained in Sec. II-B-1, the H-plane pattern (5) does not

remain valid for the back hemisphere, and, in fact, becomes a very poor

approximation beyond a 0 of about 105°. Moreover, as explained in Sec. 1

1 -A

,

the E-plane pattern in the back hemisphere, unlike the forward hemisphere,

remains accurate beyond about 105° only if the reflection coefficient is

included. Since we do not want to have to require the reflection coefficient

when using the approximate formulas (2) and (5), both the E- and H-plane patterns in the integral denominator of (14) beyond 6 = 105° were set equal to

their values at 105°. Although this particular choice for the back hemisphere

is not critical to the evaluation of gain from (2) and (5) because the power

radiated into the back hemisphere is a small fraction of the total power

radiated, some reasonable choice for the back-hemisphere pattern must be made

to evaluate the gain reasonably accurately from (14) (or more precisely, from

(14) with its integrand so modified in the back hemisphere).

The determination of gain using the approximate formulas (2) and (8) is

immediately accomplished once the constant C

Q is computed from (11) as discussed in Sec. II-B-2. The expression for gain using (2) and (8) was given in

(12b) and need not be repeated here. Gain curves computed from (12b) as well

as (14) are drawn in figure 7, which shows the considerable improvement in

gain values of both (12b) and (14) over the gain determined from the classical

Stratton-Chu results [2]. The comparison of measured and predicted values of

gain in figure 7 indicates that the formulas (12b) and (14) predict gain with

an accuracy of about 0.2 dB.

IV. CONCLUSION

This work was motivated by the need for simple, accurate expressions for

the far fields and gain of open-ended rectangular waveguide probes used in

making near-field antenna measurements. Except near the E-plane, previous

expressions based on either the Stratton-Chu formulas or first-order geometrical edge diffraction predict patterns that deviate considerably from measured

patterns--to the extent that their predicted gain values are 2 to 3 dB too

low. The alternative expressions for the far fields developed here, reduce

15
the previous limits of accuracy by about a factor of 10, over the recommended

usable bandwidth of X-band and larger standard rectangular waveguide. For Kuband and smaller standard waveguide, the wall thickness becomes an appreciable

fraction of the waveguide cross-sectional dimensions and the approximate

formulas may become less accurate.

Two methods were presented for deriving the alternative approximate formulas for the far fields (and the corresponding absolute gain). Both methods

began by assuming the sin<j> - cos<j> dependence of eq (1) with an E-plane pattern

(2) given by the classical result of integration with the Stratton-Chu formulas. The two methods differed, however, in their determination of the H- plane pattern. The first method used an E-field aperture integration (3) of

the TE 10 mode to derive the H-plane pattern (5) valid in the forward hemisphere only, whereas the second method relied upon modifying the physical

optics result (6) of the Stratton-Chu formulas with a fringe current contribution to derive the H-plane pattern (8) valid for all far-field angles. A good

estimate of the fringe current as an isotropic line source was finally

obtained through a numerical solution to the electric field integral equation

(EFIE) in Appendix I. (An alternative approach of estimating the fringe

currents by the corresponding fringe currents of the perfectly conducting half

plane gave excellent results in Appendix II for the open-ended parallel -pi ate

waveguide, but failed to produce an H-plane pattern to the desired accuracy

for the rectangular waveguide.)

In brief, the two sets of approximate formulas emerged for the far

fields— (2) and (5) or (2) and (8) inserted into (1). The former set, (2) and

(5), approximates the far fields in the forward hemisphere without requiring

the TE^q reflection coefficient for the open-ended waveguide, but predicts too narrow an H-plane pattern for angles near 90° off boresight and slightly less

reliable gain values. The latter set, (2) and (8), requires the reflection

coefficient, but yields a reliable estimate of the H-plane pattern for all

angles of the far field including the back hemisphere.

16
APPENDIX I

THE ELECTRIC FIELD INTEGRAL EQUATION (EFIE) SOLUTION

Consider the open-ended waveguide depicted in figure 1. Assume perfectly

conducting walls W, infinitesimally thin, carrying a total surface current

density K. The electric field everywhere, except within the waveguide walls,

can be expressed in terms of the vector and scalar potentials as

-4n;iaje E(r) = k2

/ K(r' )cp(r,r'

)

dS

1 0

W

+ V / (V'. K(? , )H(?,?

, )dS' , W

S (Al)

where <j>(r, r') is the scalar Green's function, e lk ^ r_r ^ / 1 r-r ' |

.

Divide K into the sum of the physical optics current K 10 of the TE 10 mode

(incident plus reflected) and the remaining "fringe current" K

f

:

K = K 10 + K

f

, (A2)

to recast (Al) into the form

-4-n:iwe E(r) = k

2

J K f

(r‘ )(j;(r,r' )dS' + V / (v ' •K f

)<|>dS' 0

W

T W

+ k

2

J K,„ 4,dS' + V J (V '.R ,UdS' (A3) u

10 w

5 10

The second line of (A3) represents the electric field produced by the truncated TE 10 mode currents, i.e., the physical optics current. They can be

conveniently expressed in terms of an integral of the fields (4) of the TE^j

mode in the infinite waveguide over only the aperture A of the mouth of the

open-ended waveguide, by means of the Stratton-Chu equations [1, Sec. 8.14]:

17
(r inside W+A)

j -4'rtiwe

() E 10

(r)

j = k

2

/ K^g c^dS ' + V / ( ) tpdS

'

(r outside W+A) I 0 ( W W

+ / [k

2

(n x H^q)^ - iwe^n x E 1Q ) x v'<|>] dS

'

A

(A4)

(As an aside, note that for r outside the waveguide, the integral over A in

(A4) is the electric field calculated from the Stratton-Chu formulas applied

to the truncated waveguide to obtain, in particular, the far fields (2) and

(6). Thus, (A4) proves that the far fields obtained by Risser [2] from the

Stratton-Chu formulas are those produced by the truncated TE^g mode currents. In other words, for r outside the waveguide, the K 10 integrals in (A3)

produce the same electric field as the Stratton-Chu formulas.) Substitute

(A4) into (A3), let the observation point r approach the surface of the

A waveguide W (from inside or outside W), take n x the resulting equation

/V ^ _ /s _ (where n is the normal to W + A into the waveguide) and set n x E and n x E-^g

equal to zero on W, to get

4uiwe n x E. (r) = n x [k

2

p

K _<pdS

1 + V p (V1 *K _ ) cpdS1 ] , (A5)

° ,nc

w

f

w

5 f

(r on W)

where E.j nc

is defined as

E. (r) = - — J [k

2 e x H 4

- iwe (e xE ) x V ' cp]dS ' , (A6)

inc 4-rti(joe . z 10 0 z 10

0 A

and p denotes the principal-value integral [16].

Equation (A5) is the desired EFIE applied to the open-ended rectangular

waveguide. It has been derived here in a form that allows the aperture TE^g

mode fields in (A6) to produce entirely the driving or incident electric field

E

1

- nc on the left of (A5). Thus the unknown surface current densities Kf in

(A5) are merely the remaining fringe currents that decay rapidly away from the

18
end of the waveguide, provided, of course, the reflection coefficient is supplied with the TE^q mode fields given explicitly in (4).

The incident field was calculated numerically from (A6) with measured

reflection coefficient r inserted into (4). The integral equation (A5) was

then solved numerically for Kf on a scientific digital computer using a moment-method numerical scheme similar to that used by Glisson and Wilton

[8]. Symmetry of the rectangular waveguide and TE^q mode, which can be

expressed easily by noting that a perfect electric and magnetic conducting

plane can be inserted at y

= 0 and x = 0, respectively, without changing the

problem [9], allowed (A5) to be rewritten in terms of fringe currents over only one quarter of the waveguide. This reduced the number of unknowns by the same factor of four. The final resulting coefficient matrix of the unknown

current Kf was inverted to find numerical values of Kf. The numerical values

of Kf were then inserted into the integrals of (A3) for r-*» in order to get

the fringe-current contribution to the far fields to be added to the StrattonChu far fields [2], which equals, as proven above, the contribution of the K 10

integrals in (A3) as r-*».

To get far field patterns over the whole forward hemisphere to the

desired few percent accuracy, this numerical EFIE solution required computer

storage larger than the available central memory of the particular scientific

computer at our disposal and a prohibitive amount of computer execution

time. However, numerical experimentation with the computer solution revealed

that the fringe currents contributing to the far-field H-plane could be

approximated surprisingly well by isotropic line sources in the y-direction at

the edges x = ±a/2. It is this result that allowed the simplified fringecurrent solution in Sec. II-B-2 of the main text.

19
APPENDIX II

FAR FIELDS AND GAIN OF OPEN-ENDED PARALLEL-PLATE WAVEGUIDE

The far fields and gain of the open-ended parallel -pi ate waveguide

(figure 1 with b+®, <j> = 0 and -n < e < n) excited by the TE 10 mode can be

found simply and accurately from the half-plane solution once the reflection

coefficient is determined (e.g., from the exact solution [10,15] or from ray

theory methods [11]). We begin with the far fields (E^, H-^) of the open-ended

parallel -pi ate waveguide approximated by the aperture integration of the twodimensional Stratton-Chu formulas with the TE-^q mode fields (4):

A E, (r*®,e) = F(r,9) e (A7a)

1 y

where

( r->® , 0

)

F(r,e)

Z

o

(A7b)

F(r£ )

E ka/2¥

o

8 [(l+r)cose + I

( l-r)

]

[-

k 9

cos(-

2

- sine)

I 9 J “ l 9 sine)

]

Hkr - l)

/kr

(A7c)

Appendix I proved that the Stratton-Chu aperture integration of the TE 10

mode fields gave the fields exterior to the waveguide and, in particular, the

far fields produced by the truncated TE^q mode currents, i.e., the far fields

of the physical optics current of the incident plus reflected TE 10 mode. Thus

the fields in (A7) neglect only the contribution from the fringe currents

(total current minus physical optics current, or equivalently, the evanescent

mode current plus the exterior surface currents). For the parallel -pi ate

20
waveguide, these fringe currents can be estimated quite accurately from the

corresponding fringe currents of the half-plane problem.

Now the remaining electric far field of the conducting half-plane, after

the far field of the physical optics current is subtracted, can be shown to be

exactly [12,13]:

E (r+«,0)

hp

. 1 a I . 1<kr -4'

-sir^ E i e

0

[cos— + cos— ] /2nkr

0 < | a | <n;

a, 0 f- ±n

( A8)

for a plane wave of amplitude Ee

y

incident on the edge and propagating at an angle a from the conducting half-plane. For the paral 1 el -pi ate waveguide,

approximate the incident field on each half plane by the two plane waves impinging on each edge, one from the incident TE^q mode and one from the

reflected TE^n mode . These two incident plane waves make an angle a with the

half plane of

a

0

= cos"

1 £

= cos-Vl-(^) 2

) (A9a)

and

u - a , (A9b)

respectively. Substituting the angles (A9) into (A8) for the two impinging

plane waves, and adding the contribution from the two half-planes forming the

open-ended paral 1 el -pi ate waveguide, gives the following approximate

expression for the electric far field excited by the fringe currents:

si E (r**>0 ) = E

cos- + r

cos— + co<

- 2

0 0

cos- + sin-^-j

,ka . , co s( —— si n© J

i

(

kr

e /2nkr

-J44

(A10)

21
The sum of (A10) and (A7a) then gives an approximate expression for the

total far field E pp

of the open-ended paral 1 el -pi ate waveguide (with the TE-^q

mode E-field parallel to the waveguide walls):

Ep

p (r-*», 0 ) = E^(r-*»,0) + E f (r+~,e) . (All)

The on-axis gain G

pp

defined with respect to an infintely extended line source

is

G

pp

Ttr | E (r+°°,o)|

PP

Z P o pp

4ixr| E | 2

PP

alE | 2 i ( 1- I

r

| 2) o k

since the input power P ppper unit length of the TE^q mode is

a f E0

1 2

|

( 1- 1 r

|

2

)

(A12a)

(A12b)

Approximate far field patterns derived from (All) (with (A7a) and (A10)

inserted), and approximate gain values from (A12), are plotted in figures 8

and 9. The comparisons contained in figures 8 and 9 with the exact solution

[10,14,15] show the significant improvement of the approximate solution (All)

(which estimates the fringe current contribution as explained above) over that

of the Stratton-Chu aperture integration (A7) (which neglects the fringe

currents)

.

22
ACKNOWLEDGMENT

Allen C. Newell, David A. Hill, and Ronald C. Wittmann of the National

Bureau of Standards (NBS), and Professor Jung-Woong Ra of the Korea Advanced

Institute of Science and Technology contributed generously to this work

through their helpful discussions and reviews. Allen C. Newell, Douglas P. Kremer, Stanley B. Kilgore, and Bill C. Yates of NBS performed and supplied

the far-field, gain, and reflection coefficient measurements of the open-ended

rectangular waveguide probes.

23
REFERENCES

[1] Stratton, J. A. Electromagnetic Theory. New York, NY: McGraw-Hill;

1941. 615 p.

[2] Risser, J. R. Waveguide and Horn Feeds, chapter 10 in Microwave

Antenna Theory and Design. Samuel Silver, ed. New York, NY: McGrawHill; 1949. 234-387.

[3] Ryan, C. E., Jr.; Rudduck, R. C. Radiation patterns of rectangular

waveguides. IEEE Trans. Antennas and Propag., Vol . AP-16: 488-489;

1968 July.

[4] Johnson, C. C. Field and wave electrodynamics. New York, NY: McGrawHill; 1965. Eq. 10.26.

[5] a) Yaghjian, A. D. Upper-bound errors in far-field antenna parameters

determined from planar near-field measurements, Part I: Analysis.

Nat. Bur. Stand. (U.S.) Tech. Note 667; 1975 October. Sec. II. A. 3.

b) Yaghjian, A. D. Efficient computation of antenna coupling and

fields within the near-field region. IEEE Trans. Antennas and Propag.,

Vol. AP-30: 113-128; 1982 January.

c) Yaghjian, A. D. Near-field antenna measurements on a cylindrical

surface: A source scattering-matrix formulation. Nat. Bur. Stand.

(U.S.) Tech. Note 696; 1977, September. Sec. 4.2.1.

[6] Kerns, D. M. General formula for voltage induced in a receiving

antenna. Digest of the national radio science meeting; 1978 November;

Boulder, CO. p. 14.

[7] Hess, D. W. Practical considerations for near-field measurements and

the use of probe correction. Digest of national radio science meeting;

1982 January; Boulder, CO. p. 186.

[8] Glisson, A. W.; Wilton, D. R. Simple and efficient numerical methods

for problems of electromagnetic radiation and scattering from sur- faces. IEEE Trans. Antennas and Propag., Vol. AP-28: 593-603; 1980

September.

[9] Tsai, L. L.; Dudley, D. G.; Wilton, D. R. Electromagnetic scattering

by a three-dimensional conducting rectangular box. J. Appl . Phys.,

Vol. 45: 4393-4400; 1974 October.

[10] Weinstein, L. A. The theory of diffraction and the factorization

method. Boulder, CO: Golem; 1969. Ch. 1.

[11] Yee, H. Y.; Felsen, L. B.; Keller, J. B. Ray theory of reflection from

the open end of a waveguide. Siam J. Appl. Math., Vol. 16: 268-300;

1968 March.

24
[12] Ufimtsev, P. Ia. Approximate computation of the diffraction of plane

electromagnetic waves by certain bodies. Part I and II. Sov. Phys.-

Tech. Phys: 1708-1718, 1957; 2386-2396, 1958.

[13] Yaghjian, A. D.; Ra, J. W. Fri nge-current diffraction coefficients for

the half plane. Unpublished Nat. Bur. Stand. (U.S.) notes; 1982

September.

[14] Heins, A. E. The radiation and transmission properties of a pair of

semi -infinite parallel plates--I. Quart. Appl . Math., Vol VI: 157-

166; 1948 July.

[15] Marcuvitz, N., ed. Waveguide handbook. New York: McGraw-Hill;

1951. Ch. 4.

[16] Yaghjian, A. D. Augmented electric and magnetic-field integral equations. Radio Science, Vol. 16: 987-1001; 1981, November - December.

25
z

Fig. 1. Geometry of open-ended rectangular waveguide.

26
Fig. 2. Measured E^ and E H power patterns of open-ended rectangular \-band

waveguide at 9.32 GHz, and comparison of measured 4

> = 45° power

pattern with 1/2(|E

E

I 2 + I

E

H

1 2

) predicted by eg (1).

27
(ap)

|(0)

3 a|

Op)

|(0)

3

a

a) X-band rectangular waveguide at 9.32 GHz

o

-5

-10

-15

-20

-25

-30

0 10 20 30 40 50 60 70 80 90

6 (degrees)

b) L-band rectangular waveguide at 1.0 GHz

Fig. 3. Amplitude and phase of E-plane field calculated from eq (2)

( ), and measured at NBS ( ).

28

(0)

(degrees)

/E

e

(

0)

(degrees)
(ap)

|(0)

H

a|

(ap)

|(0)

H

a

a) X - band rectangular waveguide at 9.32 GHz

^(degrees)

b) L-band rectangular waveguide at 1.0 GHz

Fig. 4. Amplitude and phase of H-plane field calculated from eq (5)

( ), calculated from Stratton-Chu formulas ( and measured at NBS ( ). The H-plane phase from eq (5) is

not shown because it is simply a constant phase near -90° found from

eq (9).

29

(0)

(degrees)

/E

h

(

0)

(degrees)
Frequency (GHz)

30

60

90

120

150

180

k

k

Fig. 5. Measured amplitude and phase (with respect to end) of reflection

coefficients for X-, C-, and Ku-band open-ended standard rectangular

wavegui de

.

30
a) X- band rectangular waveguide at 9.32 GHz

^(degrees)

b) L-band rectangular waveguide at 1.0 GHz

Fig. 6. Amplitude and phase of H-plane field calculated from eq (8)

( . ), calculated from Stratton-Chu formulas ( ----- ),

and measured at NBS ( ).

31
Fig. 7. On-axis gain of open-ended rectangular waveguide calculated from eq

(12b) ( ), eq (14) ( ), and the Stratton-Chu

formulas (------). The I symbols denote NBS measured values of

gain and estimated limits of error.

32
Phase

+

45

(degrees)

(Amplitude)

Fig. 8. Amplitude and phase of far field of open-ended

waveguide ( a/X = 0.75) calculated from the

( - ), the approximation (All) (. • Stratton-Chu result (A7a) (- - - - -).

parallel -plate

exact solution

. . .), and the

33
Phase

+

45

(degrees)

Gain

(dB)

Fig. 9 On-axis gain and phase of open-ended parallel -pi ate waveguide calculated from the exact solution ( ), the approximation

(All) (. . . .), and the Stratton-Chu result (A7a) (- - - -).

34
NBS-1 14A (REV. 2 -eci U.s. DEPT. OF COMM. 1. PUBLICATION OR 2. Performing Organ. Report No. 3. Publication Date BIBLIOGRAPHIC DATA

REPORT NO.

SHEET (See instructions) NBSIR 83-1689 May 1983

4. TITLE AND SUBTITLE

Approximate Formulas for the Far Fields and Gain of Open-Ended Rectangular Waveguide

5 V(riTi ur^l Y a g h j i a

n

6 . PERFORMING ORGANIZATION (If joint or other than NBS, see instruct! ons) 7 . Contract/Grant No. national bureau of standards

DEPARTMENT OF COMMERCE 8. Type of Report & Period Covered

WASHINGTON, D.C. 20234

9.

SPONSORING ORGANIZATION NAME AND COMPLETE ADDRESS (Street. City . State, ZIP)

Same as #6.

10.

SUPPLEMENTARY NOTES

_ Document describes a computer program; SF-185, FIPS Software Summary, is attached.

11.

ABSTRACT (A 200-word or less factual summary of most significant information . If document includes a significant

bibliography or literature survey, mention it here)

Approximate formulas are derived for the far field and gain of standard, openended, unflanged, rectangular waveguide probes operating within their recommended

usable bandwidth of frequencies. (Such probes are commonly used in making probecorrected near-field antenna measurements.) The formulas, which yield forward farfield power patterns and on-axis gains of X-band and larger waveguide probes to within

about 2 dB and 0.2 dB accuracy, respecti vely, assume (sin<j> - cos<}>) azimuthal angular

dependence and an E-plane pattern given by the traditional aperture integration of the

TE^g mode E- and H-fields in the Stratton-Chu equations. The H-plane pattern is estimated by two different methods. The first, and less accurate, method uses a purely E- field aperture integration to estimate the H-plane pattern. The second, more accurate, method uses an electric field integral equation ( EF IE) to show that fringe

currents near the shorter edges of the guide can be wel 1 -approximated by isotropic

line sources. The amplitude of these line sources is then determined more simply by

equating the total power radiated into free space to the net input power to the

waveguide.

12. KEY WORDS (Six to twelve entries; alphabetical order capitalize only p.roper names: m and .separate kev w ords : sr- V

edge diffraction, fringe current, physical optics, radiation pattern, rectangular

waveguide.

13. AVAILABILITY 14. NO. OF

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PRINTED PAGES

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