Paul Gemin, Tom Kupiszewski, and Arthur Radun

GE Aviation, Cincinnati, Ohio

Yan Pan, Rixin Lai, Di Zhang, Ruxi Wang, Xinhui Wu, Yan Jiang, Steve Galioto, Kiruba Haran, 

William Premerlani, Jim Bray, and Antonio Caiafa

GE Global Research, Niskayuna, New York

Architecture, Voltage and Components for a

Turboelectric Distributed Propulsion

Electric Grid (AVC-TeDP)

NASA/CR—2015-218713

July 2015
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Paul Gemin, Tom Kupiszewski, and Arthur Radun

GE Aviation, Cincinnati, Ohio

Yan Pan, Rixin Lai, Di Zhang, Ruxi Wang, Xinhui Wu, Yan Jiang, Steve Galioto, Kiruba Haran, 

William Premerlani, Jim Bray, and Antonio Caiafa

GE Global Research, Niskayuna, New York

Architecture, Voltage and Components for a

Turboelectric Distributed Propulsion

Electric Grid (AVC-TeDP)

NASA/CR—2015-218713

July 2015

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NASA/CR—2015-218713 iii

Architecture, Voltage and Components for a Turboelectric 

Distributed Propulsion Electric Grid (AVC-TeDP)

Paul Gemin, Tom Kupiszewski, and Arthur Radun

GE Aviation

Cincinnati, Ohio 45215

Yan Pan, Rixin Lai, Di Zhang, Ruxi Wang, Xinhui Wu, Yan Jiang, Steve Galioto,

Kiruba Haran, William Premerlani, Jim Bray, and Antonio Caiafa

GE Global Research

Niskayuna, New York 12309

Summary

This report contains the deliverables for the NASA Research and Technology for Aerospace 

Propulsion Systems (RTAPS) study of Architecture, Voltage and Components for a Turboelectric 

Distributed Propulsion System (TeDP).

The purpose of this effort was to advance the selection, characterization, and modeling of a 

propulsion electric grid for a TeDP system for transport aircraft. NASA is evaluating turboelectric aircraft 

propulsion to meet N+3 / N+4 goals.

Previous NASA efforts have identified blended wing body airframes as a fuel efficiency improvement 

over tube and wing airframes. Combined with a blended wing body airframe, distribution of the 

propulsion fans allows higher propulsive efficiency through reduction of drag, a higher equivalent bypass 

ratio, and boundary layer ingestion. To achieve distribution of fans beyond a limited number may become 

cumbersome with gearing and shaft arrangements, thus NASA has studied electrical propulsion 

architectures. Existing electrical propulsion systems in the required power range (sea-going ships, for 

example) would be too heavy for aerospace, either for practical integration or fuel savings. Earlier studies 

determined that distribution and protection power system components accounted for approximately 

30 percent of system mass. This study developed architecture options and circuit breaker technology 

estimating the impact on a propulsion power system.

Since the losses of turboelectric system detract from the fuel savings derived from the distributed 

concept, NASA is investigating cryogenic and superconducting electrical components. The need for 

efficient transmission of power in a cryogenic system is amplified by the additional power required to 

move any losses from cryogenic to ambient temperatures. The required cooling power is also the rationale 

behind a system where all components are cryogenic, with no interfaces to ambient temperatures, thus 

minimizing heat leak (and the power required to remove the heat). To meet these goals, the study 

proposed an architecture with hybrid circuit breakers. These breakers are estimated to provide an 

improved power density over mechanical breakers and improved efficiency over solid-state breakers.

Narrowing the range of potential system grid voltages for the electric propulsion system was also a 

major focus of the study. In selecting the voltage level, it is assumed that the high voltage system 

components will all exist in a pressurized cryogen which may be at 1 atmosphere or higher. The absence 

of electrical component exposure to low pressure gas is intended to prevent voltage breakdown. A 

supercritical cryogen is also presented as an option that would have the advantage of not allowing the 

formation of bubbles. Further, a two cryogen system (operating at two temperatures) is presented as an 

option. Such a system may allow reduced cooling power, depending on the location of losses and the 

performance and availability of cryocoolers.

Voltage selection is a balance between power conversion design, transmission cable mass, and 

cryocooler mass and power. Three parameters were varied over a wide range to determine if incorrect 
NASA/CR—2015-218713 iv

assumptions or estimates would greatly impact the preferred DC link voltage. All cases show the greatest 

decrease in the component mass, with increasing voltage, occurs by approximately 6 kV. For the 

architecture selected and the associated assumptions, a minimum occurs at about 8 to 10 kV. Given the 

wide range of estimates in cable length and power densities, a voltage of over 6 kV would capture much 

of the weight reduction; a minimum may occur between 6 kV and 10 to 12 kV. To further refine 

projections of power conversion losses, semiconductor loss testing was performed with available devices.

A dynamic model was created to allow investigation of nominal operation and limited fault scenarios. 

The philosophy of the modeling approach was to allow rapid simulation of a simplified model with 

component parameters based on projected response. Example operation of the model through controlled 

power transients and several fault cases showed stable operation of the DC power system.

For standards development related to DC power systems for TeDP aircraft, potentially relevant 

existing standards were reviewed. The marine space provides perhaps the closest analog; however, 

Medium-Voltage DC (MVDC) power systems are still not a mature technology and will require 

continued research activities. Although there are some IEEE Power Electronics Standards for Ships in 

which MVDC power systems are discussed, those standards themselves are not finalized and cannot be 

applied directly to MVDC systems targeted for aerospace propulsion application and involving the use of 

cryogenic, high power, electrical equipment. Major sections of existing standards were used to propose 

content areas for MVDC systems for aircraft.
NASA/CR—2015-218713 v

Contents

Summary......................................................................................................................................................iii

List of Acronyms and Abbreviations............................................................................................................ 1

1.0 Study Introduction............................................................................................................................... 1

2.0 Grid Architecture and Bus Voltage Range Selection.......................................................................... 3

2.1 System Requirements and Assumptions ................................................................................... 3

2.2 Architecture Trades................................................................................................................... 3

2.2.1 Baseline Architecture.................................................................................................. 3

2.2.2 Voltage Source Architecture with Fast Disconnects (Architecture 1)........................ 5

2.2.3 Modular Stack DC Architecture with Current Source (Architecture 2) ..................... 5

2.2.4 Architecture Component Breakdown ......................................................................... 9

2.2.5 Architecture Weight Assessment................................................................................ 9

2.2.6 Architecture Efficiency Assessment......................................................................... 11

2.2.7 Architecture Selection .............................................................................................. 11

2.3 Initial DC Bus Voltage Range................................................................................................. 12

2.3.1 Generator / Motor Consideration.............................................................................. 13

2.3.2 Cryogenic Electronics............................................................................................... 16

2.3.3 Cable Consideration.................................................................................................. 24

2.3.4 Power Conversion Consideration ............................................................................. 26

2.4 Conclusions............................................................................................................................. 27

3.0 Mechanical Versus Solid-State Circuit Breakers.............................................................................. 29

3.1 Comparison of Different Solutions ......................................................................................... 29

3.1.1 Mechanical Circuit Breaker...................................................................................... 29

3.1.2 Solid-State Circuit Breaker....................................................................................... 29

3.1.3 Hybrid DC Circuit Breaker....................................................................................... 30

3.2 Operation and Analysis of Proposed Hybrid Circuit Breaker................................................. 31

4.0 Design, Operation, and Weights of Hybrid Breakers and Disconnects in a TeDP Electric 

Grid ................................................................................................................................................... 33

4.1 Hybrid DC Breaker ................................................................................................................. 35

4.2 Loss, Size, and Weight............................................................................................................ 38

4.2.1 Weight and Size Estimation for the Series Inductor................................................. 38

4.2.2 Cryogenic Electromechanical Switch....................................................................... 38

4.2.3 Primary and Secondary Electronic Snubbers............................................................ 39

4.2.4 Weight and Size Estimation for Snubber Capacitor................................................. 40

4.2.5 Weight and Size Estimation for Zinc-Oxide Suppressor.......................................... 40

4.3 DC Disconnect ........................................................................................................................ 41

4.3.1 DC Contactor Preliminary Sizing Model.................................................................. 43

4.4 Hybrid AC Breaker ................................................................................................................. 43

4.5 Breaker and Disconnect Locations.......................................................................................... 44

4.6 Hybrid DC Breaker Operation in the System.......................................................................... 45

4.6.1 Motor Faults ............................................................................................................. 45

4.6.2 Ring Down................................................................................................................ 47

5.0 Transmission Cable Considerations.................................................................................................. 49

6.0 Component Sensitivities to Voltage.................................................................................................. 52

6.1 Sensitivity of Components to System Bus Voltage................................................................. 52

6.2 Sensitivity Analysis of Weight and Efficiency for Transmission Lines ................................. 52

6.3 Sensitivity Analysis of Weight and Efficiency for RTAPS Hybrid Breakers / Disconnect 

Switches .................................................................................................................................. 52

6.4 Sensitivity Analysis of Weight and Efficiency for Power Converters.................................... 53

6.4.1 Power Stack .............................................................................................................. 54
NASA/CR—2015-218713 vi

6.4.2 Passive Filter............................................................................................................. 54

6.4.3 Cooling System......................................................................................................... 55

6.4.4 Conclusion................................................................................................................ 55

6.5 Sensitivity and Scalar Summary.............................................................................................. 57

6.5.1 Generators and Motors ............................................................................................. 57

6.5.2 Breakers and Contactors........................................................................................... 57

6.5.3 Power Conversion..................................................................................................... 57

6.6 Superconducting Cables.......................................................................................................... 58

7.0 Narrowed Range of DC Voltage ....................................................................................................... 59

7.1 Approach to Narrowed Range of DC Voltages....................................................................... 59

7.2 DC Voltage Range Selection................................................................................................... 59

8.0 Final System Weight Estimate .......................................................................................................... 68

9.0 Dynamic Model and Example Case Runs......................................................................................... 71

9.1 Model Overview...................................................................................................................... 71

9.2 Component Models................................................................................................................. 73

9.2.1 Generator .................................................................................................................. 73

9.2.2 Power Converter....................................................................................................... 73

9.2.3 Generator Breaker..................................................................................................... 74

9.2.4 Contactors................................................................................................................. 74

9.2.5 Motor ........................................................................................................................ 74

9.2.6 Fan ............................................................................................................................ 75

9.2.7 Bus Voltage Controls................................................................................................ 75

9.2.8 Gas Turbine Engine .................................................................................................. 75

9.2.9 System Power Controls............................................................................................. 76

9.3 Example Cases ........................................................................................................................ 76

9.3.1 Nominal Operation with Thrust Transient................................................................ 76

9.3.2 Motor Breaker Open................................................................................................. 79

9.3.3 Motor Fault and Breaker Open................................................................................. 81

10.0 Standards Recommendation.............................................................................................................. 82

11.0 Conclusions and Recommendations.................................................................................................. 85

11.1 Conclusions and Observations ................................................................................................ 85

11.2 Recommendations................................................................................................................... 85

11.2.1 Electrical................................................................................................................... 86

11.2.2 Cryogenic.................................................................................................................. 86

11.2.3 Airframe-Propulsion................................................................................................. 86

.—Superconducting Contact ................................................................................................... 87

.—Cryogenic Semiconductor Tests and Results ..................................................................... 89

B.1 Objective ................................................................................................................................. 89

B.2 Introduction ............................................................................................................................. 89

B.3 Test Set-Up.............................................................................................................................. 90

B.4 Static Tests .............................................................................................................................. 92

B.5 Dynamic Testing ..................................................................................................................... 94

B.5.1 Resistive Test Results............................................................................................... 94

B.5.2 Inductive Test ........................................................................................................... 96

B.6 Conclusions............................................................................................................................. 98

.—Model Description ............................................................................................................ 101

References................................................................................................................................................. 105
NASA/CR—2015-218713 vii

List of Tables

Table 1.—Sizing Scenario for Architecture 1............................................................................................... 6

Table 2.—Sizing Scenario for Architecture 2............................................................................................... 8

Table 3.—Architecture Component Breakdown........................................................................................... 9

Table 4.—Component Power Density in kilowatt/kilogram......................................................................... 9

Table 5.—Cryocooler Mass Example......................................................................................................... 17

Table 6.—Comparison of Candidate DC Circuit Breaker Solutions.......................................................... 30

Table 7.—Point Design Summary .............................................................................................................. 41

Table 8.—Contactor Design Point Sizing................................................................................................... 43

Table 9.—Transmission and Distribution Mass Estimate.......................................................................... 51

Table 10.—First Order Sensitivity for Voltage Range of ~1 to ~10 kV Bus Voltage................................ 52

Table 11.—Generator and Motor Weight and Efficiency........................................................................... 57

Table 12.—Hybrid Breaker and Contactor Power Density and Efficiency ................................................ 58

Table 13.—HTSC Core Mass Per Unit Length as a Function of DC Current Rating................................. 58

Table 14.—Component Mass Summary..................................................................................................... 69

Table 15.—Cryocooler Power and Mass Estimate—Two Separate Cryo Systems Rejecting Heat 

to Ambient ................................................................................................................................ 69

Table 16.—Cryocooler Power and Mass Estimate—Single Cryogen System—Temperature Set by 

Superconductor ......................................................................................................................... 70

List of Figures

Figure 1.—Baseline Architecture (proposed in the Phase I study)............................................................... 4

Figure 2.—Possible Topology for MMC...................................................................................................... 4

Figure 3.—Architecture 1, Voltage Source Architecture, 3-Bus Multi-Feeder Architecture with 

Integrated FCL, Power Conversion, and Circuit Breaker Function............................................ 6

Figure 4.—Current Source Concept for Architecture 2................................................................................ 7

Figure 5.—Current Source Breaker-Free Architecture 2.............................................................................. 8

Figure 6.—Initial Weight Breakdown for Three Architectures.................................................................. 10

Figure 7.—Coil Supporting Structure for Air-Core Superconducting Stator Winding (Ref. 16)............... 14

Figure 8.—Air-Core Superconducting Stator Winding (Ref. 16)............................................................... 14

Figure 9.—MOSFET On-Resistance Versus Current for 77 and 300 K..................................................... 19

Figure 10.—IGBT Turn-Off as Function of Temperature .......................................................................... 19

Figure 11.—Conduction Losses as a Function of Temperature.................................................................. 20

Figure 12.—Comparison of IGBT Performance and MOSFET Performance............................................ 21

Figure 13.—Conduction Losses; MOSFET IGBT Comparison................................................................. 21

Figure 14.—Effect of Cryogenic Temperature on Maximum Power Management ................................... 22

Figure 15.—Thermal Quantities as Function of Temperature.................................................................... 22

Figure 16.—MOSFET and IGBT Losses as a Function of Temperature.................................................... 23

Figure 17.—Cryogenic Hybrid DC Breaker............................................................................................... 30

Figure 18.—Three-Bus Multi-Feeder Architecture with Integrated FCL, Power Conversion, and 

Circuit Breaker Function........................................................................................................... 33

Figure 19.—Proposed Hybrid DC Breaker................................................................................................. 35

Figure 20.—DC Disconnect........................................................................................................................ 41

Figure 21.—Current and Voltage During the Opening of Disconnect ....................................................... 42

Figure 22.—Proposed Hybrid AC Breaker................................................................................................. 44

Figure 23.—Switching Devices in Architecture 1...................................................................................... 44

Figure 24.—Circuit Condition for Motor Fault .......................................................................................... 45

Figure 25.—Typical Voltage Ringing after ZnO Suppresses Overvoltage ................................................ 47
NASA/CR—2015-218713 viii

Figure 26.—Original Hybrid DC Breaker Circuit ...................................................................................... 47

Figure 27.—DC Circuit Breaker with the Additional Resistance............................................................... 48

Figure 28.—Effective Voltage Waveform After Adding Additional Ramping Resistance........................ 48

Figure 29.—Nitrogen Cooling System ....................................................................................................... 50

Figure 30.—Neon Cooling System............................................................................................................. 50

Figure 31.—HTSC Core and Total Transmission Line Mass (700 m) vs. DC Link Voltage..................... 53

Figure 32.—Three-Level NPC Converter................................................................................................... 54

Figure 33.—Efficiency versus DC Link Voltage........................................................................................ 56

Figure 34.—Power Converter Weight versus DC Link Voltage (Note: “Cooling System” refers 

only to cryogenic pressure vessel, and not cryocoolers.).......................................................... 56

Figure 35.—Converter Power Density versus DC Link Voltage................................................................ 57

Figure 36.—Efficiency Relation................................................................................................................. 58

Figure 37.—Power Density Relation.......................................................................................................... 58

Figure 38.—Power Converter Efficiency vs. DC Link Voltage ................................................................. 60

Figure 39.—Power Converter Losses (for 100 MW) vs. DC Link Voltage ............................................... 60

Figure 40.—HTSC Core and Total Transmission Line Mass (700 m) vs. DC Link Voltage..................... 61

Figure 41.—Power Conversion System Mass Breakdown for 12.5 MW Generator Power 

Converter .................................................................................................................................. 61

Figure 42.—Power Conversion and Cryocooling and Transmission Line Mass vs. DC Link 

Voltage 3 kg/kW (input) Cryocooler scalar, 50 K cryogen, 300 K ambient............................. 63

Figure 43.—Power Conversion and Cryocooling and Transmission Line Mass vs. DC Link 

Voltage 15 kg/kW (input) Cryocooler scalar, 50 K cryogen, 300 K ambient........................... 64

Figure 44.—Power Conversion and Cryocooling and Transmission Line Mass vs. DC Link 

Voltage 0.3 kg/kW (input) Cryocooler scalar, 50 K cryogen, 300 K ambient.......................... 65

Figure 45.—Power Conversion and Cryocooling and Transmission Line Mass vs. DC Link 

Voltage 3 kg/kW (input) Cryocooler scalar, 130 K cryogen, 300 K ambient........................... 66

Figure 46.—Development of System Mass Estimates................................................................................ 68

Figure 47.—Baseline System Model in Matlab/Simulink .......................................................................... 71

Figure 48.—Baseline System Model with Major Component Labeled...................................................... 72

Figure 49.—Baseline System Model with Generator Bus Systems Labeled.............................................. 72

Figure 50.—Generator Model..................................................................................................................... 73

Figure 51.—Generator and Power Converter ............................................................................................. 74

Figure 52.—Ideal Switch Breaker Representation...................................................................................... 74

Figure 53.—Hybrid Breaker Snubber Model ............................................................................................. 75

Figure 54.—Fan Load and Fan Speed ........................................................................................................ 76

Figure 55.—Component Powers................................................................................................................. 77

Figure 56.—Engine Power and Turbine Speed........................................................................................... 77

Figure 57.—Generator and Motor Bus Voltages........................................................................................ 78

Figure 58.—System Efficiencies................................................................................................................ 78

Figure 59.—Fan Load and Speeds for a Breaker Opening on a Single Motor........................................... 79

Figure 60.—Motor and Generator Bus Voltages for a Single Motor Open................................................ 79

Figure 61.—Motor Bus Current During Fault (t = 10 s) and Breaker Open............................................... 80

Figure 62.—Hybrid Breaker Operation After Fault at t = 10 s................................................................... 80

Figure 63.—Generator Bus Voltage During Motor Fault and Breaker Open............................................. 81

Figure 64.—IGBT (left) and MOSFET (right) Full Package and Stripped Version................................... 89

Figure 65.—IGBT (left) and MOSFET (right) Full Package Overlapped With Active Silicon Area ........ 89

Figure 66.—Active Silicon Area of IGBT (right) and MOSFET (left)...................................................... 90

Figure 67.—Resistive Switching Schematic............................................................................................... 90

Figure 68.—Test Insert ............................................................................................................................... 91

Figure 69.—Test Set-Up............................................................................................................................. 91
NASA/CR—2015-218713 ix

Figure 70.—V-I Characteristic of the IGBT. Measured at LNT (solid blue), two IGBTs in parallel 

at LNT (marked as 2p IGBT), 6 IGBTs in parallel at LNT (marked as 6p IGBT). 

Characteristic at room temperature (yellow trace), and at 125 °C (red trace). ......................... 92

Figure 71.—V-I Characteristic of the MOSFET. At LNT (blue trace), at room temperature 

(yellow trace), and at 125 °C (red trace)................................................................................... 93

Figure 72.—Comparison of the V-I Characteristics of the IGBT (blue trace) and MOSFET (red 

trace). Measured at LNT........................................................................................................... 94

Figure 73.—Oscilloscope Traces for Resistive Test at 40 A...................................................................... 95

Figure 74.—First Set of Data Captured at LN2 Temperature...................................................................... 95

Figure 75.—Inductive Switching Test Set-up............................................................................................. 96

Figure 76.—Detail of the Double Pulse Test of the Inductive Switching at LNT...................................... 97

Figure 77.—Inductive Turn-On at LNT ..................................................................................................... 97

Figure 78.—Inductive Turn-Off at LNT..................................................................................................... 98

NASA/CR—2015-218713 1

List of Acronyms and Abbreviations

AMSC American Superconductor 

BIU Bus Interface Unit

BSCCO Bi2Sr2Ca2Cu3O10

COP Coefficient of Performance

CORC Conductor On Round Core

DUT Device Under Test

EMI Electromagnetic Interference

FCL Fault Current Limiter

GaN gallium nitride

HVDC high-voltage DC

LN2 liquid nitrogen

MgB2 magnesium diboride

MMC Modular Multilevel Converter

MSDC Modular Stack DC

MTBSI Mean-Time-Between-Service-Interruption

MVDC medium-voltage DC

N2 nitrogen

Ne neo

NPC Neutral-Point-Clamped

SCS Surround Copper Stabilizer

Si silicon

SiC silicon carbide

SSCB Solid-State Circuit Breaker

TeDP Turboelectric Distribution Propulsion

THD Total Harmonic Distortion

YBCO yttrium boron copper oxide

ZnO zinc oxide

1.0 Study Introduction

The purpose of this effort is to advance the selection, characterization, and modeling of a propulsion 

electric grid for a TeDP system for transport aircraft. NASA is evaluating turboelectric aircraft propulsion 

to meet N+3 / N+4 goals.

Previous NASA efforts have identified blended wing body airframes as a fuel efficiency improvement 

over tube and wing airframes. Combined with a blended wing body airframe, distribution of the 

propulsion fans allows higher propulsive efficiency through reduction of drag, a higher equivalent bypass 

ratio, and boundary layer ingestion. To achieve distribution of fans beyond a limited number may become 

cumbersome with gearing and shaft arrangements, thus NASA has considered electrical propulsion 

architectures. Existing electrical propulsion systems in the required power range (sea-going ships, for 

example) would be too heavy for aerospace, either for practical integration or fuel savings. Additionally, 

the losses of such systems would detract from the fuel savings derived from the distributed concept. For 

the improved power density and component efficiency, NASA is investigating cryogenic and 

superconducting electrical components.
NASA/CR—2015-218713 2

This study had seven primary milestones:

1. Range of bus voltage and grid architecture selection for TeDP,

2. Mechanical versus solid-state circuit breakers comparison for cryogenic operation

3. Mass and efficiency model development for all grid components,

4. The development of sensitivities to bus voltage for all components,

5. The selection of a narrowed range of DC voltage,

6. Dynamic state-variable model development, and

7. Standards development recommendations

This report addresses these milestones and shows the progress of the work. For example, component 

mass and efficiency estimates were made early in the study to select architecture concepts, but were later 

refined as the architecture detail was developed. The report covers the early study estimates, as well as the 

estimates reached by end of the study.

NASA efforts provide several high-level assumptions to use for evaluation of the systems. Any of 

these assumptions were open to modification if a rationale was discovered. These assumptions are listed 

below.

• A turboelectric system

• 50 MW generation system

• 50 MW fan loads

• 1 to 10 kV power system

• All cryogenic system—no components at ambient temperatures

• Rotating machines were expected to be fully superconducting, synchronous machines

Narrowing of the voltage range was a major focus of this work. Several other areas were out-ofscope. Specifically, the cryogenics system that supports the cold components was presumed to be 

sufficiently redundant and was not part of this study; linking of the propulsive electrical grid to any other 

aircraft electrical systems was not part of this study; and detailed modeling of the superconducting 

machines was not part of this study.

The sections below describe the study approach and results to reach the defined milestones.
NASA/CR—2015-218713 3

2.0 Grid Architecture and Bus Voltage Range Selection

2.1 System Requirements and Assumptions

Per NASA guidance, the bus voltage range to start with was 1 to 10 kV. The high power parts of the 

electrical system are expected to be entirely cryogenic, from the generator stator windings to the motor 

stator windings, without any high-current electrical leads to room temperatures. Also per scope definition, 

thermal modeling was not a significant part of the tasks; the cryogenic system that supports the cold 

components was presumed to be sufficiently redundant and was not part of this study. Detailed machine 

design was not part of the study. The rotating machines were expected to be fully-superconducting, 

wound-rotor, synchronous machines with superconducting windings on both the rotors and the stators.

The N3-X TeDP Baseline architecture includes two wingtip-mounted turboshaft engines, and each 

drives two cryogenically cooled superconducting electrical generators. Power electronics convert the 

variable frequency AC power to DC so as to improve distribution efficiency and eliminate the 

requirements to sync the power sources. NASA specifies 50 MW of total system power, with 25 MW in 

the event of a single engine failure. Therefore each power plant (one engine / two generators / converters) 

is sized to provide 25 MW, the minimum power required at take-off for the single engine failure scenario 

as recommended by NASA studies (Ref. 1). Two power plants’ total power level is 50 MW. Fail-safe 

requirements from (Ref. 1) indicates that no single point failure should yield catastrophic consequences at 

the aircraft level. Redundancy design and component sizing in this study will follow this rule so that with 

any single point failure in the electric grid, the remaining system can provide 25 MW propulsion power.

2.2 Architecture Trades

A recent NASA study (Ref. 1) discussed promising architectures. However, it also demonstrated that 

circuit breakers could constitute 30 percent of the weight of the entire TeDP system, which was a 

component not previously included in earlier studies. Two new architectures will be proposed 

incorporating a power rating increase from 22.37 to 25 MW per NASA instruction and targeting advanced 

techniques to limit the usage of circuit breakers while maintaining high reliability. It needs to be noted 

that this stage of the study was completed with the assumption of 50 MW generation capability, but only 

25 MW fan load. When the rating of elements in this section are listed as a percentage, this is the 

percentage of 25 MW, the required power for rolling take-off. The final estimates and architecture use 

50 MW generation and 50 MW load component ratings.

2.2.1 Baseline Architecture

This architecture is based on Architecture 3 developed in the referenced NASA study (Ref. 1), which 

allows the same rating for all four generators and three buses. Redundant feeders provide power rerouting 

capability during a bus / generator / engine failure. The following study will take it as the Baseline 

Architecture for comparison. For an easy and clear comparison, the Baseline Architecture is shown in this 

report again in Figure 1.
NASA/CR—2015-218713 4

Figure 1.—Baseline Architecture (proposed in the Phase I study)

Figure 2.—Possible Topology for MMC
NASA/CR—2015-218713 5

A newly developed Modular Multilevel Converter (MMC) technology (Refs. 2, 3, and 4) can be 

applied to provide integrated power conversion, fault current limitation, and fault current isolation. 

Figure 2 demonstrates a possible topology of MMC in a HVDC application. In our study, only one or two 

modules are needed due to the much lower voltage level. The adopting of MMC can bring the Baseline 

Architecture to a much more compact design where a single MMC can replace the fault current limiter, 

rectifier, and circuit breaker. However, MMC technology relies on the use of large capacitors, which 

greatly detracts from the weight benefits of removing circuit breakers, so much so that this technology is 

not taken into consideration for any of the architectures.

2.2.2 Voltage Source Architecture with Fast Disconnects (Architecture 1)

The reduction of circuit breakers in Architecture 1 is achieved by a new fault isolation scheme with 

the usage of super-fast disconnects as illustrated in Figure 3. A disconnect is a kind of mechanical contact 

which does not have current interrupting capability, but can open the circuit while no current flows in it. It 

is much lighter than a circuit breaker, about 600 kW/kg compared to 200 kW/kg, as the study in 

Appendix A shows. The modification made in Architecture 1 is replacing most of the circuit breakers 

with disconnects to reduce the system weight while maintaining high availability.

In Architecture 1, there are four DC circuit breakers at the starting end of wing transmission lines to 

interrupt the current from generators when a fault occurs in the downstream system. A DC circuit breaker 

is also put near each motor converter to prevent back feeding from motors during an upstream fault. Other 

circuit breakers originally designed to be in the Baseline Architecture are replaced by super-fast 

disconnects, such as the circuit breakers at the bus end of the wing transmission lines and the circuit 

breakers at both ends of the central feeders. If a fault occurs in the distribution system (on the wing

transmission, the bus, or the feeder), the isolation scheme is that the related circuit breakers will rapidly 

open to stop feeding the fault from the generator and motors, then DC disconnects can operate to isolate 

the faulty section, followed by close of circuit breakers to recover the power feeding from the generator to 

the remaining healthy system. With quick response of circuit breakers and super-fast disconnects, the 

whole process of fault detection, isolation, and recovery can be done in milliseconds with only small 

disturbances imposed on the system. Some study was done to examine the availability of superconducting 

disconnects and its performance. The finding is summarized in Appendix A.

Table 1 shows the sizing scenario for Architecture 1. Engines, generators, and AC/DC converters are 

sized by the one engine failure scenario according to NASA studies. The existence of redundant feeders, 

which increases the connectivity between bus and propulsor, allows the propulsors to be optimally sized for 

their own redundancy. Given the large number of propulsors, they are sized such that 14 out of total 16 

propulsors can provide the full take-off power. Outer buses are sized the same as their connected generators. 

The inner bus is sized slightly larger to supply eight propulsors connected to it through primary feeders. 

While primary feeders are sized to the same rating as propulsors, redundant feeders are sized to spread the 

remaining power from its connected bus to propulsors that are primarily supplied from the other bus.

2.2.3 Modular Stack DC Architecture with Current Source (Architecture 2)

An alternative architecture to reduce weights of circuit breakers is to apply the Modular Stack DC 

(MSDC) technology developed by GE (sponsored by RPSEA), which connects all the generators and 

loads in series and uses a by-pass switch to short out the faulted branch (Refs. 5, 6, 7, and 8). The 

concept of MSDC is shown in Figure 4. Basically it is a current source architecture where generator 

converters run in a current control mode. Generators and motors are connected in series and a ring bus is 

formed. The bidirectional Bus Interface Units (BIU) interface generators and motors to the bus and 

provide the bidirectional power flow capability. For line-line faults occurring on any of the branches, 

instead of isolating the faulted branch using a circuit breaker, a bypass switch in parallel with the branch 

will be closed to bypass the faulted branch. Therefore, no DC circuit breakers are needed on branches, 

while lighter bypass switches and disconnects can do the job of fault isolation.
NASA/CR—2015-218713 6

Figure 3.—Architecture 1, Voltage Source Architecture, 3-Bus Multi-Feeder Architecture with Integrated FCL, Power 

Conversion, and Circuit Breaker Function

TABLE 1.—SIZING SCENARIO FOR ARCHITECTURE 1

Component Sizing method

Engine 1 engine failure

Generator/converter 1 engine failure (100%/2)

Transmissions lines 1 engine failure (100%/2)

Bus

Outer bus 1 engine failure (100%/(3-1))

Inner bus 8 propulsors (100%/(16-2) * 8)

Feeder Primary feeder 1 propulsors (100%/14)

Redundant feeder 1 engine failure (50% – 7.14% * 4)/4)

Propulsor (drive/motor/fan) 2 propulsor failure (100%/14)
NASA/CR—2015-218713 7

Figure 4.—Current Source Concept for Architecture 2

Table 2 indicates the scenario that sizes each element in the system. The size of each element is 

indicated in terms of percentage of 25 MW, the required power for rolling take-off. As is outlined in the 

table, a single engine failure scenario dominates all other off-nominal operations for this architecture.

Just like the concept of multiple buses in Architecture 1, when adopting MSDC more than one ring 

bus will be used to provide redundancy and improve system reliability. Figure 5 shows the system layout 

with this concept applied. The architecture consists of four subsystems. Each subsystem consists of one 

generator, one AC/DC converter, one bus with an associated energy storage device, and four propulsors 

with AC/DC converters as motor drives. All of the AC/DC converter and DC/AC converters are 

connected to the ring bus via a BIU. The BIU can control the power flow between the converter and the 

ring bus which is carrying constant current.

In normal operation, the load can be distributed among the 16 propulsors, e.g. each generator feeds 

25 percent of the system power to each ring bus. During the one engine failure scenario, the two normally 

open switches between the two middle ring buses will be closed. And the two normally close switches in 

the two ring buses will open. Thus, the two middle ring buses forms a new ring bus. Via this new ring 

bus, one of the generators powered by the healthy engine can keep feeding power for the middle 

8 propulsors. In other words, 12 propulsors can continue working together to provide the required power 

for rolling take-off.

If a line-to-line short circuit fault occurs in the system, such fault can be cleared by the BIUs and bypass switch. For example, if feeder cable is shorted, the AC breaker in the propulsor will open to cut 

energy fed from the fan. At the same time, the by-pass mechanism in the BIU will also close to stop 

energy from the ring bus. After the feeder is bypassed by the BIU, the normally open switch will close to 

fully isolate the fault. Since in the ring bus system all of the generator and loads are connected in series 

and the operation of BIU and switch will not interrupt the current in the ring bus, the other propulsors will 

not even sense such fault. If the ring bus is shorted, the ring bus needs to be shut down by closing all of 

the BIU connected to the ring bus. In the worst case, (a permanent short circuit fault occurs inside the ring 

bus), the whole ring bus will be shut down, which equals to one engine failure fault.
NASA/CR—2015-218713 8

TABLE 2.—SIZING SCENARIO FOR ARCHITECTURE 2

Component Sizing method

Engine 1 engine failure

Generator/converter 1 engine failure (4 or 8 propulsors)

Transmission lines 1 engine failure (4 or 8 propulsors)

Bus

Outer bus 1 engine failure (4 propulsors)

Inner bus 1 engineer (8 propulsors)

Feeder 1 propulsors (100%/14)

Propulsor (drive/motor/fan) 1 engine failure (100%/12)

Figure 5.—Current Source Breaker-Free Architecture 2

If a line-to-ground fault occurs in the system, a fault current may flow through the ground path. In 

this case, the ring bus connected with the faulty point needs to be shut down by the BIU, following a 

similar method to the one mentioned above. Part of the system connected to the ring bus can be recovered 

after the fault is cleared by operating the corresponding disconnects.

Compared with the voltage source architectures, no DC breaker is used here. All of the fault can be 

cleared by BIU and by-pass switches. This is the key reason why the weights of circuit breakers can be 

reduced in the ring architecture.
NASA/CR—2015-218713 9

2.2.4 Architecture Component Breakdown

Table 3 breaks down the proposed two architectures and the Baseline Architecture in terms of the 

number and power rating of each component.

2.2.5 Architecture Weight Assessment

The weight of two architectures is assessed and compared against the Baseline Architecture. The 

weight model for individual components will be updated in the following report sections; in this early 

study phase the component density values and assumption adopted in the earlier NASA study were used 

(Ref. 1). The calculated power density values in kilowatt/kilogram are summarized in Table 4 (Ref. 1).

Figure 6 shows the weight breakdown for three architectures. Compared with the Baseline 

Architecture, Architecture 1 reduces the system weight by about 370 kg, while Architecture 2 reduces by 

160 kg. And the weight reduction is mainly from the protection represented by the yellow and green bars. 

As discussed earlier, Architecture 1 uses lighter fast disconnects to replace many of the circuit breakers, 

while Architecture 2 adopts a breaker-free architecture to reduce the protection weight.

TABLE 3.—ARCHITECTURE COMPONENT BREAKDOWN

Component Baseline Arch 1-VSI Arch 2-CSI

Count Rating Count Rating Count Rating

Machines

Generators 4 50% 4 50% 4 2-33%, 2-67%

Motors 14 8.33% 16 7.14% 16 8.33%

Converter

AC/DC 4 50% 4 50% 4 2-33%, 2-67%

DC/AC 14 8.33% 16 7.14% 16 8.33%

Protection

AC CBs

4 50% 4 50% 4 2-33%, 2-67%

14 8.33% 16 7.14% 16 8.33%

DC CBs

8 50% 4 50%

56 28-8.33%, 28-5.56% 16 7.14%

Disconnects

4 50% 4 2-33%, 2-67%

64 32-7.14%, 32-5.36% 16 8.33%

BIUs 4 2-33%, 2-67%

16 8.33%

Cable

Transmission 4 50%, 2-30 m, 2-40 m 4 50%, (2-30 m, 2-40 m) 4 2-67%, 30 m; 2-33%, 40 m

Feeder 28 14-8.33%, 14-5.56%, 5 m 32 16-7.14%, 6-5.36%, 5 m 16 8.33%, 5 m

Bus 3 50%, 2-5 m, 1-10 m 3 50%, 2-5 m, 1-10 m 4 2-67%, 5 m; 2-33%, 5 m

TABLE 4.—COMPONENT POWER DENSITY

IN KILOWATT/KILOGRAM

Component Power density 

(kW/kg)

Generators 78.5

Motors 47.1

Converters 40

DC circuit breakers 200

AC circuit breakers 350

BIUs 200

Disconnects 600

Cables 500 A/kg/m
NASA/CR—2015-218713 10

Figure 6.—Initial Weight Breakdown for Three Architectures

The weight of Architecture 1 is smaller than that of Architecture 2. This is because redundant feeders 

in Architecture 1 provide power rerouting capability for propulsors during a bus / generator / engine 

failure. This allows a reduction in propulsor size. This power rerouting capability is hard to implement in 

Architecture 2 with ring buses, especially during a bus failure.

Figure 6 indicates that more than 50 percent of the system mass comes from AC/DC and DC/AC 

converters. Considering the preferred DC grid architecture of the TeDP electrical system, it may be worth 

investigating DC electric machine technologies. In this way, it may be possible to complete DC 

architectures without the use of power electronics and possibly achieve a significantly lower system mass.

One option is the common DC machine comprised of a DC stator field winding, rotating armature 

coil, rotating commutator, and stationary brushes. Challenges exist with the design of a high voltage, 

high-speed commutator. In this machine, an AC voltage is generated within the rotating armature winding 

and the AC voltage is rectified by the rotating commutator and stationary brushes. Conventional DC 

machines generally operate at DC voltages below 1 kV because the solid conductor commutator 

segments, which lie next to each other on a cylindrical rotor, must be electrically insulated from each 

other. Electrical arcing between the commutator segments will be more difficult to prevent at higher 

voltages.

2.2.5.1 Generator Option

Another option is the DC homopolar machine, which utilizes the Lorentz force between a constant 

radial unidirectional magnetic field and an axial current flowing within the winding on a rotor. This type 

of machine does not require a commutator because no alternating current or voltage is produced within 

the machine. However, brushes are still required to transfer the current from the rotor to a stationary wire.

In addition, this type of machine is inherently a high-current, low voltage machine since the rotor winding 

is essentially a single turn coil. Thus it may be difficult to produce a brush to carry the high current 

necessary for this application. Two examples of this type of machine have been built (Refs. 9 and 10).
NASA/CR—2015-218713 11

2.2.6 Architecture Efficiency Assessment

The first order efficiency assessment will be performed assuming consistent efficiency for 

components (generator, motor, and power converter) in different architectures, and it is assumed that 

there is no loss on superconducting cables and contactors. Then the assessment is about how many power 

conversion stages the power will flow through from generators to motors. For the baseline study, every 

unit power will flow through a total of five power conversion stages: generator side AC/DC converter, 

one transmission line solid-state circuit breaker, two feeder solid-state circuit breakers, and motor side 

DC/AC drive. For Architecture 1, there are a total of three power conversion stages since two feeder 

solid-state circuit breakers are replaced by hybrid breakers with superconducting contactors. For 

Architecture 2, power converters include: generator side AC/DC rectifier, generator BIU, motor BIU, and 

motor side DC/AC drive (a total of four power conversion stages with Architecture 2). With the above, 

the efficiency of Architecture 1 will be better than Architecture 2. Both them are better than the baseline.

2.2.7 Architecture Selection

The weight comparisons above estimate both Architecture 1 (voltage source architecture with fast 

disconnects) and Architecture 2 (Modular Stack DC Architecture with Current Source) were lighter than the 

Baseline Architecture, and that Architecture 1 is 210 kg lighter than that of Architecture 2. However, the 

relatively small estimated weight delta also requires that other metrics such as the bus voltage, flexibility, 

scalability, and technology readiness are considered to select one architecture for the following study.

In Architecture 1, the wing transmission, buses, and central distribution all operate at the same 

voltage level. While in Architecture 2 generator branch and motor branches are connected in series, wing 

transmission and bus voltage will be the sum of the connected motor branch voltages. Therefore, with the 

same voltage level of the motor converter, Architecture 2 will be with a higher bus and wing transmission 

voltage level. The insulation requirements will increase. As discussed in Section 2.3 and specifically 

Section 2.3.4, we propose to set the motor converter voltage to the maximum available blocking voltage 

of a single semiconductor device to optimize overall system efficiency. That voltage is estimated to be at 

least 5 kV based on certain projected assumptions. Then the bus voltage in Architecture 2 will be that 

voltage times the number of motor branches connected in a ring. When increasing the number of motors 

(propulsors) to scale up the system, a concern will be whether the high bus voltage reaches a value where 

the insulation becomes extremely challenging. (With the increase of voltage level, the design and control 

of electric field distribution under DC may become more challenging within both cables and cable joints 

and terminations. Due to the volume effect of insulation breakdown stress, moving towards higher voltage 

may require the increase of insulation, which can affect the overall size and weight of the system.)

Another advantage associated with voltage source type of Architecture 1 is actually demonstrated in 

the earlier NASA study. It is easy to increase the connectivity between generators and motors in a voltage 

source architecture to reduce the total power rating of propulsors. And considering the short distance of 

distribution cables required in the aircraft architecture, the trade of redundant feeders / transmission for 

reduced rating of propulsors usually brings in overall system weight benefit. This power rerouting 

capability is hard to implement in the ring architecture because of the series connection structure and the 

need for the interface unit, BIU.

A ring architecture has some attractive features for today’s MV grid and it was adopted in GE’s 

recent subsea program. First, it removes the need for a MVDC circuit breaker, which is a challenge with 

today’s technology. Second, industry application usually prefers high transmission voltage (about 160 kV 

in GE’s subsea program for example) to reduce the cable conduction losses, and the interface unit (BIU) 

in those applications can also be used to step down the voltage to a suitable level the load prefers. This 

function needs to be implemented by another stage of power converter in the voltage source architecture.

So the BIUs are not additional components for the subsea application. However, with the assumption that 

the system under study is to be entirely cryogenic, we do not necessarily need to increase the transmission 
NASA/CR—2015-218713 12

voltage to reduce the thermal loss, and the entire system (including generation, transmission, and load) 

can run at the same voltage. Therefore, BIU is just an additional level of power conversion used to 

facilitate the ring bus structure and it washes out the “breaker free” feature of the ring architecture.

Due to the above reasons, the voltage source architecture (Architecture 1), shown in Figure 3, is 

recommended to use for this study. However, if some basic assumptions can be changed, the conclusions 

could be reversed, leading to some very interesting research topics for the future study. For example, if 

the DC motor can be used in the system instead of the synchronous machine, the DC/AC converter and 

AC/DC converter can both be replaced by simple DC/DC converters. The related benefits may not be 

obvious for the voltage source architecture, since the DC breaker is still needed to block the energy stored 

in the system’s DC capacitors. However, this can bring much more benefits for the ring architecture, since 

the function of DC/DC converter can be integrated into the BIU. In other words, the weight and loss 

related to the DC beaker can be truly eliminated in the ring architecture. In addition, if a true bipolar 

force-commutated device becomes available, both the weight and loss in the DC breaker and BIU can be 

reduced significantly. This could also change the conclusion.

2.3 Initial DC Bus Voltage Range

Bus voltage is a critical parameter in future aircraft DC distribution systems which have a significant 

impact on system weight. Generally speaking for conventional conductors, higher voltage levels are 

desired as power demand increases. Increasing the voltage level allows the conductor current to decrease 

for the same power rating, which leads to smaller conductor dimension and lowers conductor weight.

However, higher voltage levels require thicker insulation which contributes to an increase in system 

weight. With the assumption that the system under study is to be entirely cryogenic, the thermal loss / 

ohmic (I2

R) of superconducting cables and machines are negligible, a different thinking should be taken.

Maximum power transmitted at high efficiency can be achieved by operating the system at an optimal 

voltage such that all anticipated currents through the superconducting cables are close to, but less than, 

the critical current (within safe margin), and the voltage losses in the dielectric are small. Furthermore, for 

aircraft application, cable insulation and electrical system layout must be designed appropriately 

considering Paschen’s Law in order to avoid breakdown and partial discharge, especially in a cryogenic 

environment with lower air pressures. Given construction options with cryogenic cables and components, 

it may be desirable to operate the cryogenic system at pressures above ambient. It is expected that 

pressure structures will contribute more greatly to system weight than dielectrics materials within the 

pressurized environment. Due to the complications summarized above, the relationship between bus 

voltage and system weight is not expected to be monotonic (Ref. 1).

DC bus voltage selection needs a coordinated consideration of generator voltage and propulsor 

voltage. Voltage gain of power converters, which determines converter topology selection, depends on the 

voltage ratio of generator voltage over bus voltage or the ratio of bus voltage over propulsor voltage.

Matching DC bus voltage and operating current appropriately with generator / propulsor voltage / current 

can minimize power conversion and simplify converter structure. It can, therefore, potentially reduce 

system weight and improve power efficiency. In addition, DC bus voltage level and generator / propulsor 

voltage level will determine voltage and current ratings of converter components, including 

semiconductor devices and passive filtering components.

DC bus voltage selection cannot be separated from DC system architecture design. Different system 

architectures may involve different power conversion technologies and different protection schemes, thus 

different switching, isolating, and current-limiting components.

Another important factor is the type of semiconductor devices that will be used in the system. It will 

be desired to understand the current density of different device technologies (like MOSFET, IGBT, 

IGCT, GTC and Thyristor) at different voltage levels. As shown in many studies (Refs. 11, 12, and 13), 

the electrical characteristics of the semiconductor devices change (sometimes drastically) when operated 
NASA/CR—2015-218713 13

at cryogenic temperature. The changes in both dynamic and static behavior are due to changes in physical 

parameters such as charge mobility, lifetime, etc. These changes depend on the operating temperature as 

well as the device internal architecture and their operating mode. GE is leading the adoption of silicon 

carbide technology, but since silicon carbide devices will not work well under a cryogenic environment 

and devices like silicon are well suitable for cryogenic temperature, those devices will be the main focus 

in the study. With the above-mentioned aspects, system architecture, converter technology, 

semiconductor devices, protection scheme, generator voltage, and propulsor voltage are all parts of the 

optimal system voltage selection equation.

Passive components such as capacitors and inductors also need to be optimized for operation in a 

cryogenic environment. Generally speaking the voltage breakdown strength of capacitors decreases, but 

their capacitance seems quite stable at low temperature (Ref. 13). Common inductor materials’ core losses 

increase in a cryogenic application; therefore, it is best to avoid them (Refs. 14 and 15).

The manufacturability of the equipment also plays a major role in determining the optimal system 

voltage. Studies can show the theoretical optimal system voltage, but ultimately each piece of equipment 

must be producible. When studying the optimal system voltage, manufacturability metrics should be 

included so that the ability to produce the equipment is taken into consideration when optimizing (Ref. 1).

Overall, given the total power requirement of 50 MW, a systematic trade-off study is done in this task to 

obtain an initial range of DC bus voltage roughly within the 1 to 10 kV range. The study incorporates grid 

architecture design, converter topology selection, generator / motor voltage consideration, cryogenic 

electronics characteristics, superconducting cable characteristics, and power converter voltage consideration.

2.3.1 Generator / Motor Consideration

The entire electrical system is assumed to be superconducting and cryogenic and the electric 

generators and propulsion motors are assumed to be fully-superconducting, synchronous machines, 

including the wound rotor and stator windings (Ref. 15).

Generally superconducting electrical machines are built with an air-gap stator winding; in other 

words, iron teeth are not utilized between the coil sides of the armature winding. This is because the iron 

teeth would easily saturate given the very high magnetic field levels produced by the superconducting 

field winding. The coils are often still supported and insulated from each other using non-magnetic 

“teeth.” However, there may be other ways to support the coils without using non-magnetic “teeth.” An 

example coil supporting structure is shown in Figure 7 (Ref. 16). A stator tooth is represented by 

item 370, a stator coil is represented by item 380, and the machine frame is item 360. The stator yoke or 

back-iron, which is depicted just radially outward of the coils, may or may not be laminated magnetic 

material. Figure 8 (Ref. 16) shows an image of an air-gap winding in which item 110 refers to the 

armature winding placed within the air-gap between the stator yoke 180 and the rotor 220. A magnetic 

back-iron 180 is not strictly necessary, but it may be utilized to contain the amount of flux that fringes 

into the frame of the machine, thereby minimizing eddy current loss within the frame.

Several factors must be considered when selecting the voltage rating of the electric machines in the 

TeDP system, which is directly translated to the system’s DC-bus voltage by a scaling factor. These factors 

include electric machine power density, efficiency, resilience to fault conditions such as short circuits, and 

system stability. The goal of the electric machine design for the TeDP system is to maximize the power 

density and efficiency of the electric machines, while maintaining acceptable short circuit and system 

stability conditions. However, the following fundamental analysis will provide rationale for why the voltage 

ratings of the electric machines have only a secondary effect on these critical system factors. The medium 

voltage range is very typical for conventional electric machines in the 1 to 10 MW range. Although 

insulation requirements will increase as the voltage increases, there is no evidence suggesting that the 

electric machine’s coil construction or design will be limited by the assumed voltage range of this study.
NASA/CR—2015-218713 14

Figure 7.—Coil Supporting Structure for Air-Core 

Superconducting Stator Winding (Ref. 16)

Figure 8.—Air-Core Superconducting Stator Winding (Ref. 16)

2.3.1.1 Power Density

In order to maximize the power density (1) of an electric machine at a given speed, one must 

maximize the shear stress (τ) within the air gap of the machine. Since shear stress (2) is directly 

proportional to the average air gap flux density (β) and armature current loading (σ), one should 

maximize both values within the cooling and current carrying capabilities of the SC wire utilized in the 

rotor and stator windings. Note that current loading refers to the amp-turns divided by the circumferential 

distance around the air-gap, and current density refers to the amount of amp-turns divided by the crosssectional area of pure conductor within a coil.

The air gap flux density (3) and current loading (4) are proportional to the current density in the field 

winding and the armature winding, respectively. Air gap flux density is primarily driven by the amount of 

flux produced by the field winding. Therefore, it is assumed that the field and armature windings will 

operate at the current density and AC loss limits of the future state-of-the-art superconducting wire in 

order to maximize the power density of the electric machines.

As in any conventional electric machine, the armature phase voltage (5) is simply proportional to the 

rate of change of the air-gap flux times the number of turns per phase. If the number of turns per phase is 
NASA/CR—2015-218713 15

increased in order to increase the rated voltage of the machine, then the current per turn and the conductor 

area per turn must also decrease in order to keep the current density constant within the armature winding.

The opposite is true if the voltage is decreased by decreasing the number of turns. In this way, the 

terminal voltage of the machine can be exchanged with the line current of the machine by changing the 

number of turns per phase. In addition, a similar exchange can be performed by changing the number of 

parallel circuits within the machine. Both methods of adjusting the rating of the machine do not affect the 

power density of the machine on a first order basis.

A secondary effect of the voltage rating on the power density of the electric machine is related to the 

amount of electrical insulation that is necessary between the machine’s turns and phase windings. As the 

voltage increases, the amount of insulation increases, which decreases the amount of space available for 

current conducting wire. Thus the current loading may decrease. However, the decrease in current loading 

is much less than proportional to the increase in voltage. Therefore, the effect on power density is 

minimal as the voltage increases.

This study will use a baseline generator plus cooler power density of 34.6 kW/kg and a motor plus 

cooler power density of 20.9 kW/kg based on the mass estimates calculated in Reference 15 assuming a 

hypothetical BSCCO material for the superconducting wire.

= ωmτ

VOL

S 2 (1)

τ = σβ (2)

DL

Np

π

/

β = O (3)

D

k2NIaNp σ = (4)

1

2 O

k

fN VLLπ / = (5)

S electric machine apparent power (VA)

VOL air gap volume (m3

)

τ shear stress (psi)

ωm angular speed of the rotor (rad/s)

σ current loading (A/m)

β average air gap flux density (T)

Np number of poles

∅ flux per pole

D air gap diameter

L air gap length

N number of turns in armature circuit

Ia armature current

f frequency

VLL line-to-line armature voltage

k1 winding scaling factor

k2 winding scaling factor
NASA/CR—2015-218713 16

2.3.1.2 Efficiency

As stated in previous studies, the efficiency of the electric machines within the TeDP system will 

affect the weight of the cryogenic cooling components required, and, therefore, the power density of the 

overall system. As is the case with power density, the voltage of the superconducting electric machine has 

only a secondary effect on the efficiency of the machine.

The electrical loss within an electric machine can be thought of being driven by the same two “fields” 

that produce output power from the machine. In other words, electrical loss is dependent on current 

loading (or current density) and flux density.

AC loss within the superconducting wire is defined in Reference 15 as a combination of eddy current, 

hysteresis, and coupling eddy current losses. Note that all the loss components of the AC loss are 

dependent on current density and flux density. Eddy current and hysteresis loss within other magnetic 

structures within the machine are also driven by the flux density level. A conventional electric machine 

will possess the bulk of these losses within the iron stator and rotor cores. The modified Steinmetz 

Equation (6) is an example of the relationship between the loss in an iron material and the flux density 

level experienced by the iron material.

2B2 P K f B K f LossCore = h + e α β (6)

PLoss Core core (laminated iron) loss

Kh hysteresis loss coefficient

Ke eddy current loss coefficient

f frequency

B magnetic flux density

As stated previously, the voltage of an electrical machine can be varied by changing the number of 

turns without changing the flux density or current density levels within the machine. Therefore, the losses 

related to the flux density and current density levels of the machine can be isolated from the voltage of the 

machine.

This study will use a baseline generator plus cooler efficiency of 99.55 percent (roughly 99.95 percent

motor only efficiency), and a motor plus cooler efficiency of 99.97 percent (roughly 99.997 percent motor 

only efficiency) based on the efficiency estimates calculated in Reference 15, assuming a hypothetical 

BSCCO material for the superconducting wire.

2.3.2 Cryogenic Electronics

Cryogenic Power Electronics is a definition that refers to power electronics equipment operated at 

very low temperatures. While the most common explored operative temperature is 77 K because it is 

quite easy to reach and maintain, the largest gain in both efficiency and size reduction may be achieved at 

higher temperatures, especially if the weight and efficiency of the cryocooler needs to be accounted for.

Additionally, different materials have different optimal working temperatures (where optimal is intended 

as the temperature where they exhibit the lowest losses); therefore, an optimal system may be a one with 

subcomponents operated at different temperatures. This report lightly touches on the main power 

electronics components, highlighting benefits and downsides of their operation at cryogenic temperatures.

Table 5 provides an example of the system weight impact of both component inefficiency and cryogenic 

temperature. This report does not consider operative temperatures that are not cryogenic even if it is 

suggested that a more extensive study, including a comparison with a high temperature system, can be 

done in the future.
NASA/CR—2015-218713 17

TABLE 5.—CRYOCOOLER MASS EXAMPLE

Cryocooler Mass vs. Efficiency

(30% Carnot, 3 kg/kW input)

50 MW Loss

(kW)

Mass at 77 K 

(kg)

Mass at 100 K 

(kg)

99.95% 25 724 500

99.90% 50 1,448 1,000

99.50% 250 7,240 5,000

99.00% 500 14,481 10,000

95.00% 2,500 72,403 50,000

90.00% 5,000 144,805 100,000

2.3.2.1 Capacitors and Insulators

The behavior at cryogenic temperatures of capacitors strongly depends on the material chosen as 

dielectric. Polypropylene, polycarbonate, and mica exhibit very good behavior at cryogenic temperature 

(same capacitance and smaller dissipation factor) while tantalum exhibits an increase in dissipation factor 

(Ref. 13). Other material (Ref. 17) exhibits a much larger breakdown voltage. It is also suggested the use 

of cryogenic liquids as a good material for cryogenic insulators (Ref. 18).

2.3.2.2 Inductors

The behavior at cryogenic temperatures of magnetic materials strongly depends on the magnetic 

material considered. In published work (Ref. 14) it has been found that for some materials permeability 

decreases as temperature decreases (for example ferrite and Kool Mu), others remain constant (for 

example MPP and High Flux), while the material resistance decreases as the temperature decreases 

(leading to increase in eddy currents induced losses) for all the material considered. The windings of the 

inductor will produce much less losses; therefore, the total losses of an inductor in cryogenic environment 

are comparable with total losses of the same inductor at room temperature.

Inductor for electromagnetic interference (EMI) shielding becomes much more effective at cryogenic 

temperature (given the fact that the magnetic material is more lossy at cryogenic).

It must be noted that the materials considered in literature are designed for room temperature and 

above; a magnetic material for optimized performance at cryogenic temperatures can be designed. So, it 

can realistically be predicted that in the near future, given the need, such a material can be designed and 

high efficiency cryogenic inductors could be produced.

2.3.2.3 Semiconductors

This section presents an overview of the two most popular semiconductor materials, and the two most 

common devices structures and their expected behavior in a cryogenic system. The two materials 

considered are silicon carbide (SiC) and silicon (Si). Other materials that may be used to manufacture 

switching devices are not considered in this report because it may have fallen in disuse (such as 

germanium), is not currently considered for high power application (such as GaN), or is too early to be 

considered (such as diamond). Heterojunctions are also not considered.

2.3.2.3.1 Silicon Carbide (SiC)

Silicon carbide material is a very promising material for high performance switches in power 

electronics. This material, due to its large band-gap and higher maximum operating temperature, exhibits 

an excellent switching behavior and quite low conduction losses, making it desirable for both high 

frequency and high power applications. In addition, this technology is currently entering the market with 

commercially-available product grade components.
NASA/CR—2015-218713 18

Given the high performance behavior of this material, it has been considered for this application but 

ultimately rejected for cryogenic applications.

The performances of this material degrade as the operative temperature falls. This degradation has 

been observed for two structures, Schottky diodes (Ref. 19) and MOSFETs (Ref. 20), and, without going 

into details, it can be attributed to the increase of the trapped-electron density that occurs when the 

material temperature decreases.

SiC have been observed to display a drop in breakdown voltage capability as the temperature 

decreases.

2.3.2.3.2 Silicon (Si)

Silicon is the most common material used for fabrication of power electronics switches. It is used for 

both high frequency applications and high power applications. This material is so commonly used for its 

availability, its properties, and the available know-how that has been created after many years of its use.

While at room temperature and above it does not perform as advantageously as SiC material does, 

especially at medium power high frequency applications, it still remains the material of choice for high 

power applications, and any other cost sensitive applications. It has been shown in numerous works (as 

the latest review shows (Ref. 21)) that the behavior of this material greatly improves, in terms of lower 

losses (both conduction and switching) and switching speed, as the operative temperature decreases. The 

improvements are not monotonically related to the operative temperature: depending on the structure and 

the doping level, the devices built out of silicon exhibit an optimal temperature that could be between 70 

and 200 K.

Si have been observed to display a drop in breakdown voltage capability as the temperature 

decreases. The breakdown capability drops to about 70 percent of its nominal value (at room temperature) 

when the operative temperature is about 77 K (Refs. 11 and 22).

2.3.2.3.2.1 MOSFET

The MOSFET is generally a majority carrier device, which means that the conduction is associated 

with a resistive-like behavior, and the charge flow interruption is not associated with carrier lifetime.

Because the mobility of the electrons increases as the temperature decreases, in this structure the 

conduction losses greatly decreases as the temperature decreases.

Figure 9 from Reference 12 shows the great improvement (in terms of much smaller on-resistance) 

when the device is operated at low temperatures.

The switching losses and speed are not affected as much as the conduction losses, at least for a 

traditional MOSFET structure.

It has been reported that a cool-MOS structure exhibits large advantages for both conduction losses 

(Ref. 22) and switching losses (Ref. 23).

2.3.2.3.2.2 IGBT

The IGBT structure is mostly minority carrier device structure. This structure is designed to conduct 

large currents in a reduced area, making this the device of choice when high power density is required. 

The conduction is associated to both carrier mobility and carrier lifetime, and its switching behavior is 

associated mostly with carrier lifetime.

As the temperature decreases the mobility increases—making the “resistive-like” conduction losses 

decrease—but the lifetime decreases—making the conduction losses associated with carrier density 

increase. As these two phenomena are in opposition, this device structure exhibits an optimal working 

temperature where the mobility is high and the lifetime is not too low (Ref. 11).
NASA/CR—2015-218713 19

Figure 9.—MOSFET On-Resistance Versus Current for 77 and 300 K

Figure 10.—IGBT Turn-Off as Function of Temperature

As the switching behavior benefits from a shorter lifetime, the switching time and switching losses 

decrease as the temperature decrease, making possible the use of this device at high frequency and high 

power.

Figure 10 shows the faster switching time (Ref. 24). It also shows that there is an optimal temperature 

at which the device is the fastest; above or below this temperature the device performances degrade. 

Figure 11 shows that a similar behavior applies for conduction losses. The device exhibits an optimal 

temperature. It must be noted that different IGBTs exhibit different optimal temperatures and different 

gains—this is an indication that the performances are connected with doping levels and device design. An 

IGBT device can be designed to have optimal performances at cryogenic temperatures.
NASA/CR—2015-218713 20

Figure 11.—Conduction Losses as a Function of Temperature

2.3.2.3.2.3 Comparison

In order to select the proper component as MOSFET or IGBT structure, a comparison of losses and 

cooling must be done.

MOSFETs behave as a resistor; therefore, if they are paralleled, not only the losses per device 

decrease, but the system efficiency improves. This is not true for the IGBTs—as the conduction losses are 

proportional to the current that flows in them, paralleling multiple devices does not improve the system 

efficiency; it only reduces the power loss per device. This is true at any temperature. For this comparison 

only, structures with same active areas are considered.

Figure 12 shows a comparison of IGBT and MOSFET performances where the choice of the proper 

structure depends on the cooling removal capability. Figure 12(a) illustrates that with the choice for the 

cooling capability system represented, the MOSFET can manage more power than the IGBT for the same 

amount of losses; while Figure 12(b) shows the opposite—if the system can remove larger losses, than the 

better choice will be the IGBT.

Figure 13 shows a comparison between conduction losses of IGBT and MOSFET. The data for the 

IGBT have been measured, while the data for the MOSFET have been extrapolated from previous works.

It must be noted one more time that, if massive paralleling is considered, then the MOSFET structure 

will always be the choice. Massive paralleling will bring additional issues such as signal coordination, 

device selections, layouts, etc. that may make this approach unrealistic.
NASA/CR—2015-218713 21

(a) MOSFET Better Performing (b) IGBT Better Performing

Figure 12.—Comparison of IGBT Performance and MOSFET Performance

Figure 13.—Conduction Losses; MOSFET IGBT Comparison

When operating at cryogenic temperatures, the maximum power that can be removed from the 

junction increases. This is due to the fact that the thermal conduction of silicon increases as the 

temperature drops, making it easier for the heat to reach the cryo-bath, therefore, limiting the junction 

temperature rise for same heat produced. This allows for larger maximum currents that the device can 

carry (as shown in Figure 14).

Figure 15 shows the calculated thermal conductance and specific heat for silicon as a function of 

temperature according to Reference 25.

As mentioned in the previous section, an IGBT structure gains even during switching performances. 

As the temperature drops, the lifetime of carriers decreases making the switching transition much faster, 

therefore, allowing for higher operational frequencies and higher power capabilities.
NASA/CR—2015-218713 22

Figure 14.—Effect of Cryogenic 

Temperature on Maximum 

Power Management

Figure 15.—Thermal Quantities as Function of Temperature

In general, it is possible to state that as the temperature is decreased, a MOSFET structure exhibits the 

most improvement during conduction, while the IGBT structure exhibits less improvement during 

conduction, but a much improved transition time and transition losses.

Figure 16 represents in general terms the effect of lowering the temperature on the two structures: at 

room temperature the MOSFET has low switching losses and high conduction losses, the IGBT has high 

switching losses and low conduction losses. As the temperature decreases the switching losses of the 

MOSFET decrease a little, but conduction losses decrease a lot. The opposite is true for the IGBT.
NASA/CR—2015-218713 23

Figure 16.—MOSFET and IGBT Losses as a Function of Temperature

2.3.2.3.3 Doping Effects

The doping level has a large effect on the optimal operational temperature of the device as well as the 

overall performances. As the temperature decreases and carrier lifetime decreases, the concentration of 

ionized dopant also decreases, culminating in carrier freeze-out where there are no more charges available 

for conduction. This phenomenon is prevalent in minority carrier devices. In order to counter-balance this 

effect, a device with larger doping level may be more beneficial, especially at deep cryogenic levels.

2.3.2.3.4 Semiconductor Selection Conclusions

In conclusion, for cryogenic applications at high power and high voltage, when massive paralleling is 

not considered, the semiconductor device of choice is the Si IGBT optimized for low temperature 

operations (it is difficult to compare with coolMOS given the fact that only two works have been found—

one does not cover transition times and losses (Ref. 22), and the other one (Ref. 23) is dated). As the 

device will be able to carry much more current than it can at room temperature but it will be able to block 

less voltage (about 30 percent less) than it could at room temperature, it is advisable that the topology of 

choice will take advantage of these new characteristics: high current capability, lower voltage capability, 

and higher switching frequency capability. The cryogenic temperature of choice will be addressed in the 

following sections.

2.3.2.4 Cryocoolers

Cryocooling is the most important reason that kept and still keeps cryogenic power electronics from 

entering the market. Most of the works on this topic focus on the benefits introduced by the 

semiconductor devices, some focus on the benefits and behaviors of the passives, and very few on the 

physics of cryocooling and the power balance involved at system level.

If the cryo-bath is kept at 77 K and the heat to be extracted is rejected at 300 K (room temperature), 

the Carnot cycle Coefficient Of Performance (COP) (not to be confused with efficiency) is only 

34 percent. Considering that today’s practical coolers exhibit a COP of about 30 percent of the Carnot 

cycle, it is clear that a realistic COP is about 10 percent, which means that for 1 W of heat to be extracted 

at 77 K and rejected at 300 K the system will consume 10 W, making the total losses 11 W. The losses to 

be compared with a system operating at room temperature are 11 W and not 1 W. As described, it makes 

it very difficult to build a cryogenic power electronics system with a clear advantage over a traditional 

system. For example, with 2.1 percent loss as the target, the power converters at room temperature need 

to be designed with 97.9 percent efficiency, while the power converters at cryo-temperature need to be 

designed with 99.8 percent efficiency (0.2 percent loss) since 2 percent power (10X) is needed to reject 

the heat to room temperature.
NASA/CR—2015-218713 24

Because of the nature of the Carnot cycle and cryocoolers, the COP drastically increases as the 

cryogenic temperature increases; for example, if the cryo-bath is at 150 K and rejects at 300 K, the COP 

of the Carnot cycle is 100 percent and assuming same percentage of the Carnot cycle COP of 30 percent

(even if it improves for a smaller variation of temperatures), it brings the overall COP to 30 percent. In 

this new example about 3.5 W will be needed to remove 1 W, bringing the total losses to 4.5 W: a much 

smaller number than the first case.

A breakeven plot of required efficiency of a device operating at 150K versus its efficiency at room 

temperature is reported in Reference 26. For example, if room temperature device efficiency is 

80 percent, and it is cooled down with a Gifford-McMahon cycle, its minimum efficiency needs to be 

about 95 percent; if it is cooled with Stirling cycle it needs to be 90 percent.

2.3.2.5 Power Electrics Conclusions (Device, Temperature, and Voltage)

This short study did not take into consideration the use of SiC material at room temperatures or 

above. While such a study will be very helpful, and the authors wish it could be done in the near future, it 

is outside the scope of this work. Additional complications introduced by such a study will be the losses 

in the connectors’ cryo-to-room-temperatures in terms of conduction losses and added losses rejected into 

the cryogenic bath, and the losses of the new cooling system to keep the SiC within an acceptable 

temperature.

The use of cryogenic environment will not only allow power savings in the single elements, but will 

also allow the overall system size reduction by the use of small passives (such as capacitors and 

inductors) by enabling much higher switching frequencies. Additionally semiconductor devices, as well 

as passive elements, have higher reliability and are more robust to over currents when operated at 

cryogenic temperatures than at room temperature.

In conclusion, for the semiconductor devices, it is recommended the use of Si IGBT optimized for a 

high cryogenic temperature (i.e., 150 K). The IGBT should be rated for about 5 kV at cryogenic 

temperatures (see study on topology efficiency) and kAs level. While such devices cannot be found today, 

it is quite realistic extrapolating that they will exist in the near future. (There are currently devices rated 

for 6.5 kV at room temperature and devices rated for several kA at temperatures larger than 300 K.)

2.3.3 Cable Consideration

Superconducting wire has changed the industry with its revolutionary ability to conduct over 100 

times the electrical current of conventional wire. The high power density of superconducting wire 

dramatically reduces the size, weight, and overall cost of large-scale electrical systems, and also increases 

the efficiency of the electrical systems compared with systems based on traditional copper wire.

HTS cables use tapes or wires made of superconducting materials as current carrying elements.

Bi2Sr2Ca2Cu3O10 (BSCCO) with a critical temperature of 110 K and YBa2Cu3O7 (YBCO) with a critical 

temperature of 92K are commercially available superconductors used in HTS cable.

BSCCO as the first generation HTS wire has been produced in multi-filamentary structure in silver.

BSCCO wire is available in km length with a current density of 100 A/mm2 and above at 77 K.

YBCO as the second generation of HTS wire has been produced mainly in the form of a thin, wide, 

multilayer tape. YBCO tape is currently available in hundreds of meters length with a current density of 

100 A/mm2 at 77 K. YBCO wire is fabricated by an automated, continuous process using thin film 

deposition techniques, similar to those used in the semiconductor industry. Therefore, YBCO has 

inherently better price / performance ratio than BSCCO.

Alternatively, magnesium diboride (MgB2) with a critical temperature of 39 K is used for 

superconducting cables operated at lower temperatures. MgB2 wire has been produced in multifilamentary structure in matrix. MgB2 wire is available in km length, with a current density approaching 

1000 A/mm2 at 20 K.
NASA/CR—2015-218713 25

SuperPower and American Supercondutor (AMSC) are the two major manufacturers of produce HTS 

wire.

For SuperPower wire, the total wire thickness is only 0.095 mm. It uses Surround Copper Stabilizer 

(SCS) with no sharp corners, which is good for high voltage applications and can be easily soldered for 

joints. It uses high-strength Hastelloy substrate for better mechanical properties and lower AC losses.

SuperPower is able to deliver 2g HTs wire in 100 to 300 m typical length with high critical current of 100

A (standard) in 4 mm width tape. At 77 K, this wire can operate up to 1 T maximum field.

SuperPower 2G HTS wire has been successfully constructed and tested in various applications, such 

as transmission cable, Fault Current Limiter (FCL), high filed magnet, and coils. For example, Advanced 

Conductor LLC constructed a Conductor On Round Core (CORC) cable consisting of 20 tapes, wound 

into six layers, forming a cable with an outer diameter of 7 mm and then wound into a two layer, 12 turn 

magnet. The CORC cable magnet reached a winding current of 1950 A at 20 T. A record winding current 

of 4100 A at 19.81 T was demonstrated in 2012. Another example is a 30 m 2 G HTS cable constructed 

by Sumitomo Electric Industries and installed in National Grid systems in 2008 as part of the Albany 

HTS cable demonstration. AC losses of 0.36 W/kA-m were demonstrated.

In superconducting power cable, layers of HTS tapes, which transmit the power, are stranded around 

a metal of hollow former. The HTS tapes are surrounded by high voltage insulating material, commonly 

referred as dielectric. Furthermore, a cryostat insulates the whole cable core against ambient temperature.

The HTS tapes are usually cooled with liquid nitrogen (LN2) (77 K), which is pumped through the cable 

cryostat. Consequently, the dielectric is also immersed in LN2. Therefore, this cable is generally called a 

cold dielectric cable. Such a cold dielectric cable may be of benefit to the aerospace example considered, 

as it allows conductor elements to see pressures higher than ambient.

In a HVDC cable design, the inner layers of HTS tapes transmit power while the outer layers are 

grounded. Induced current in outer layer will eliminate the electromagnetic fields of the inner layers.

Therefore, the electromagnetic field is contained inside the superconducting screen. The cable inductance 

can be reduced significantly.

For medium voltage (up to 72 kV), concentric cable offers the most compact design with the best 

utilization of HTS tapes. All three phases and a screen, each separated through dielectric, are stranded 

around a hollow former. The former can be used as a return channel for LN2. Concentric cables exhibit no 

external magnetic fields as well.

The design of superconducting HVDC cable is very similar to the design of superconducting HVAC 

cable. The inner HTC layers are separated through the dielectric from a screen consisting of copper wire 

only. With bipolar HVDC cable system, with voltage levels up to ±320 kV a transmission power of 

4.5 GW and even more is easily attainable.

From nkt cables’ HTS Triax energy cable systems datasheet, their three phase MVAC cable 

constructed with 1G BSCCO-type HTS tape can provide phase current up to 3.4 kA, and phase voltage up 

to 42 kV. The weight of the HTS cable core (three phases, thermal insulation not included) is ranging 

from 3.5 kg/m (1.25 kA phase current) to 8 kg/m (3.4 kA phase current).

In this study, each wing transmission line needs to carry power of 12.5 MW and current of 

6.25/V kA, while V is the DC bus voltage (in kV) in the bipolar DC system. Lower voltage level will lead 

to higher current rating for superconducting cables. Therefore, DC bus voltage selection should also 

consider superconducting cable’s current carry capability. At the same time, higher insulation requirement 

is associated with higher bus voltage. The impact of voltage on cables is more of a trade-off between 

cable rating and insulation requirement. Figure 6 illustrates the early study estimate that distribution 

cables only contributed around 5 percent of the whole system mass; however, when the cable insulation is 

considered in addition to the superconducting core, and if greater lengths are required due to limitations in 

bend radius, then cable mass will contribute to the optimum system voltage. The cable impact will favor 
NASA/CR—2015-218713 26

higher voltage within the considered range of approximately 1 to 10 kV. Section 6.6 covers the mass 

assumptions used for both the superconducting core and the transferline.

2.3.4 Power Conversion Consideration

As mentioned in Section 2.3.2, at cryogenic temperature the equivalent resistivity of both IGBT and 

MOSFET will be significantly reduced and IGBT will have a much lower resistivity compared with 

MOSFET. Thus, even considering the constant voltage drop across the IGBT, the total voltage drop of 

IGBT is still much less than that of MOSFET for a given silicon area at a high level current.

To keep high power density, IGBT is selected to build the cryogenic converter.

For a converter, the power rating can be estimated as

P NV I sys = dc (7)

in which, Psys is the converter power rating, Vdc is the single device blocking voltage, N is the number of 

series connected devices in the converter to block the total DC bus voltage, and I is the system current 

flowing through each device. N is related to converter topology, e.g., N is 1 for a 2 level converter and is 2 

for 3 level converter topologies.

For an IGBT device, the conduction loss in the device can be expressed as

(V R I)I V I

P V I

ceon

loss cond ce

0 0

_

= + ≈

= (8)

in which, Ploss_cond is the device conduction loss, Vce is the voltage drop of the IGBT when it is carrying 

current, V0 is the constant voltage drop part and Rceon is the equivalent resistor of the IGBT. Since the 

constant voltage drop is dominant, Vce can be estimated close enough as V0. Cryogenic test conducted at 

GE-GRC shows that with about six Si IGBT chips in parallel, the conduction loss on the equivalent 

resistor of the IGBT is about one tenth of the loss on the constant voltage drop. This supports the 

assumption of using the constant voltage drop to estimate the conduction loss. V0 is determined by the 

silicon material and semiconductor manufacture process and does not change much with device voltage or 

current rating.

The converter loss mainly consists of two parts: device conduction loss and switching loss. If the 

percentage of device conduction in the overall converter loss is k, the converter efficiency can be 

expressed as

dc

dc

loss

loss cond

kV

V

kNV I

NV I

kP

P

0

0

_

1

1

1

= −

= −

η = −

(9)

Because k and V0 are constants, higher Vdc is desirable to achieve higher efficiency. In other words, IGBT 

with higher blocking capability can help increasing system efficiency.

For example, if the target converter system efficiency (η) is higher than 99.8 percent, the voltage drop 

across the single IGBT is 1V, and conduction loss is 80 percent of the overall converter loss, then
NASA/CR—2015-218713 27

V kV

V

V

kV

V

dc

dc

dc

5

2 2 2

0.2%

80%

1

2 2 2

0.2%

99.8%

0

≥

× × ≤

× × ≤

η ≥

(10)

In this calculation, the first factor of 2 represents the two stages of power conversion: one on the 

generator side and on the propulsors side. The second factor of 2 represents the two converters in each 

stage: the DC/AC or AC/DC converter and the DC breaker for voltage source architecture or the BIU for 

ring architecture. The third factor of 2 represents the number of series connected device in each converter 

during the operation.

The calculation is only an example to show the idea. The exact number may change with different 

projected V0 and k. The voltage threshold also depends on how technology evolves. If DC motor 

technology is mature enough to be used in 2035, for the ring architecture, the function of DC/AC and 

AC/DC converter can be integrated into the BIU, so the impact of the second factor of 2 can be 

eliminated. In addition, if bipolar IGBTs become available, the third factor of 2 can also be eliminated.

Thus, to meet the same efficiency number, devices with lower blocking voltage can be used. Or in other 

words, the system efficiency can be improved with the same device.

For the voltage source architecture, the transmission line voltage can be equal to the single device 

blocking voltage (Vdc), if a simple two level converter is used. In this example case, it can be 5 kV. For 

the ring architecture, the transmission line voltage is MVdc, M is the number of propulsors.

2.4 Conclusions

One voltage source architecture (Architecture 1) and one current source ring bus architecture 

(Architecture 2) are proposed and compared against the Baseline Architecture (Architecture 3 proposed in 

first study).

• Both architectures reduce system weight. Weight reduction in Architecture 1 (8.6 percent) is 

achieved by an advanced fault isolation scheme with the usage of high power density fast 

disconnects. Architecture 2 adopts a breaker-free architecture to reduce the system weight (by 

3.6 percent).

• Architecture 1 uses redundant feeders to reduce the propulsor size, which makes it 210 kg 

(5 percent) lighter than Architecture 2. This power rerouting capability is hard to implement in 

Architecture 2.

• In Architecture 1 the whole system (including wing transmission, buses, and distribution feeders) 

operate at the same voltage level, while in Architecture 2 bus voltage is multiple times of feeder 

voltage. A higher insulation requirement is associated with Architecture 2.

• The need for a Bidirectional Interface Unit in Architecture 2 detracts from its weight benefit 

achieved by removing the need for DC circuit breakers.

• Based on the above reasons, Architecture 1 is selected for the following study. The development 

of new technologies, such as DC machines and bipolar semiconductor devices, will possibly 

change the selection and significantly reduce the system weight.
NASA/CR—2015-218713 28

Given the initial range of 1 to 10 kV, DC bus voltage study incorporates grid architecture design, 

generator / motor voltage consideration, cryogenic electronics characteristics, superconducting cable 

characteristics and power converter voltage consideration.

• Machine weight mainly depends on its power rating and it is not sensitive to voltage rating (see

Section 2.3.1.1).

• A 1 to 10 kV medium voltage range is very typical for conventional electric machines in the 1 to 

10 MW range. Although insulation requirements will increase as the voltage increases, there is no 

evidence suggesting that the electric machine’s coil construction or design will be limited by the 

assumed voltage range of this study.

• For cryogenic applications of high power, when massive paralleling is not considered, the 

semiconductor device of choice is the Si IGBT optimized for low temperature operations.

• Cryogenic device will be able to carry much more current than it can at room temperature and 

block less voltage (30 percent less) than it could at room temperature.

• The impact of voltage on cables is a trade-off between cable rating and insulation requirement.

When only considering the superconducting core mass for cables, one may conclude that they 

would be only a secondary effect on the system weight; however, cable construction for cryogen 

and thermal insulation significantly add to the weight and will impact voltage level selection, as 

shown in later sections.

• Semiconductors with the maximum blocking voltage should be used to increase system 

efficiency. At cryogenic temperature, the equivalent resistor of IGBT device is negligible and the 

forward voltage drop demonstrates a constant value and dominates IGBT’s conduction loss. To 

achieve a certain target converter efficiency level, a lower bound of DC voltage can be calculated.

A calculation based on Architecture 1 and the best approximation of related coefficients indicates 

DC bus voltage should be greater than 5 kV to meet 99.8 percent converter efficiency 

requirement.
NASA/CR—2015-218713 29

3.0 Mechanical Versus Solid-State Circuit Breakers

3.1 Comparison of Different Solutions

DC circuit breakers add system weight in the TeDP Electrical System. The first phase study shows 

that approximately 30 percent of the entire electrical system weight is from DC circuit breakers (Ref. 1).

The voltage source architecture proposed above uses lighter fast disconnects to replace many of the DC 

circuit breakers, which significantly reduce the weight of DC circuit breakers to 10 percent of the system 

weight. In this task, we review and compare different types of DC circuit breakers and propose a solution 

for high-efficiency and high power density DC breakers.

The three promising approaches we consider in this study include:

1. Conventional DC breakers used at ambient temperature

2. Cryogenic Solid-State Circuit Breaker

3. Cryogenic Hybrid Circuit Breaker

The steady-state rating of the DC breakers in this study is considered to be 1 kA continuous in a 

10 kV circuit.

3.1.1 Mechanical Circuit Breaker

Cryogenic DC circuit breaker technology has not been developed. For electromechanical DC circuit 

breakers, the cryogenic temperatures would cause the material to become brittle and fragile (Ref. 1).

Therefore, conventional DC breakers have to be used at ambient temperature in the TeDP application, and 

connections are needed through the cryostat wall.

Conventional DC breakers have been nominally designed for approximately 4 kA and up to several 

kilovolts (Ref. 27). They are able to break up to 50 kA of fault current (Refs. 27 and 28). The power 

density is in the neighborhood of 100 kW/kg, with contact resistance in the range of 10 to 100 μΩ 

(Ref. 29). The advantage of using conventional DC breakers is that it is a well-known technology, with a 

proven track-record. So, it is low risk. However, the large size and high response time (up to 10 to 

100 msec) makes it not suitable for architecture we proposed in the selected Architecture 1. Meanwhile, it 

can only be used at ambient temperature. The need for connections through the cryostat wall increases 

system complexity.

3.1.2 Solid-State Circuit Breaker

Compared with mechanical circuit breaker, Solid-State Circuit-Breakers (SSCB) have been proposed 

for the grid applications (Ref. 30). Solid-state circuit breakers based on high power semiconductors 

potentially offer enormous advantages when compared to the conventional mechanical solutions, since 

solid-state breaker is able to switch in a few microseconds. Hence, the maximum current will never be far 

away from the rated current and the whole process will just last for around a few hundreds of 

microseconds. Meanwhile, advanced semiconductor packaging techniques can enable adequate cooling 

performance. However, compared with mechanical breaker, higher on resistance of the semiconductor 

switch also creates high conduction loss. Based on the previous analysis and summary of cryogenic 

semiconductors in this report, the silicon IGBT is preferred in a high power cryogenic application. With 

the cryogenic technology, the on resistance of the semiconductor can be greatly reduced. Nevertheless, 

the forward voltage drop of the silicon IGBT generates high loss for conducting the main current, which 

is still a concern in the TeDP system because of the high efficiency target. Therefore, a hybrid solution 

that combined both mechanical and solid-state technologies is desired for the DC protection in the 

aviation system.
NASA/CR—2015-218713 30

3.1.3 Hybrid DC Circuit Breaker

Literature (Ref. 31) describes a hybrid HVDC breaker. During normal operation, the current only 

flows through the lower-impedance bypass, which consists a fast mechanical disconnect in series with a 

semiconductor-based load commutation switch. When a fault occurs, the load commutation switch 

immediately commutates the current to the main HVDC breaker, which is separated into several sections 

with individual arrester banks and the fast disconnect opens (Ref. 31).

The recent advances in cryogenic contactors (Ref. 32) suggest that the eventual development of 

cryogenic hybrid DC breakers is likely, because a cryogenic contactor can be used to construct the 

cryogenic hybrid breaker shown in Figure 17.

The breaker is comprised of a cryogenic contactor in parallel with an electronic snubber. During 

normal operation, the main current only goes through the cryogenic contactor, which leads to very small 

conduction loss. The parallel semiconductor only conducts the current during transient process. Because 

DC voltage level in the TeDP electrical system is up to 10 kV, probably only two switching sections are 

needed in the snubber path considering single device’s capability around 5 kV. Then the voltage across 

the contactor will be around 2 V (around 0.7 V forward voltage drop for single silicon IGBT device). 

Detailed analysis shows that the contactor itself can open to force commutation without generating arc or 

damaging itself. Therefore, the load commutation switch in series with the contactor can be removed to 

further reduce the conduction loss. The detailed analysis of this cryogenic hybrid DC breaker will be 

shown in the next section.

Based on the preliminary comparison, the characteristics of the three DC protection candidates are 

summarized in Table 6. With the consideration of fast response, higher efficiency, compact size, and 

lighter weight, the hybrid circuit breaker combined with cryogenic technology is selected as the DC 

protection system for the TeDP electrical system.

Hybrid DC breaker

Cryogenic Contactor

Electronic Snubber

Figure 17.—Cryogenic Hybrid DC Breaker

TABLE 6.—COMPARISON OF CANDIDATE DC CIRCUIT BREAKER SOLUTIONS

Response time Fault current Arcing Size and weight Efficiency

Mechanical circuit breaker 10 to 100 ms High, poorly controlled Yes Large (100 kW/kg) High

Solid-state circuit breaker 10 to 100 μs Controlled Minimal Medium (200 kW/kg) Low

Hybrid circuit breaker 100 μs to 1 ms Slightly higher than fault current Minimal Medium (175 kW/kg) High
NASA/CR—2015-218713 31

3.2 Operation and Analysis of Proposed Hybrid Circuit Breaker

Operation during a fault is as follows: When the fault is detected by whatever protection / control 

logic is employed by the power system architecture, the electronic snubber is placed in the conducting 

state and the cryogenic contactor is opened. When the contacts part, the current will commutate into the 

snubber. Commutation will be detected by a rise in voltage across the snubber, due to the conduction of 

the snubber electronics. At that point, the snubber will be turned off, and the voltage across it will rise in 

response to the inductive energy stored in the rest of the network. The snubber will incorporate zinc oxide 

(ZnO) voltage suppressors to clamp the voltage and to absorb inductive energy trapped in the network.

After the fault is cleared, the breaker is reset as follows: The snubber is switched to the conducting 

state, and then the contactor is closed.

There are several parameters to be considered in this application:

1. Steady-state current of 1 kA and open circuit voltage of 10 kV.

2. Contact resistance. On the order of 1 μΩ is projected for a cryogenic contactor, resulting in 

steady-state losses of 1 W per breaker.

3. Contactor operation time. Using a repulsion coil, on the order of 100 μsec should be possible.

4. Interruption current. During a fault, the current through the DC breaker will rise. The rate of rise 

can be limited by including a superconducting inductor in the circuit.

5. Snubber conduction voltage. It is projected that this will be on the order of 2 V for electronic 

devices capable of conducting 1 kA for 100 μsec, and capable of withstanding 10 kV stead-state, 

and pulse voltages of 16 kV for a few milliseconds.

6. Contactor-snubber loop inductance. On the order of a 100 nH is expected.

These parameters are related as follows. When the contactor contacts part, the current that they were 

carrying will commutate into the electronic snubber. This will create a brief pulse of voltage across the 

contacts. Details of the voltage waveform depend on stray capacitance, loop inductance, and the effective 

resistance of the snubber. As the contacts part, their resistance will go from 1 μΩ to infinity over a very 

short time period, typically 10 to 100 nsec. During this time, the voltage will rise from 1 mV, to several 

volts, then back down to the steady-state voltage of the snubber, on the order of 2 V. It can be shown that 

the amount of energy dissipated in the contacts during the commutation process is approximately equal to 

the energy stored in the loop inductance. We expect that the current during the commutation process with 

be on the order of 2 kA, so with a 100 nH loop inductance, we expect to dissipate 0.2 J in the contacts.

With this small value we do not expect any damage to the contacts.

It is projected that the ratings of cryogenic contactors could easily reach 50 kA. None-the-less, we 

would like to minimize the amount of current flowing through the contacts at the moment of interruption, 

in order to minimize the voltage spike when the contacts part, and to minimize the amount of energy 

dissipated in the contacts. To that end, we propose including a superconducting inductor in the circuit to 

limit the rate of rise of current during the fault. An inductance of around 1 mH is suggested. That will 

limit the rate of rise of current to 10 A/μsec during a fault, so the current through the contacts will be 

around 2 kA when the contacts part 100 msec after fault initiation. Of course, the ZnO voltage clamp in 

the electronic snubber will have to be sized to absorb this energy, approximately 2000 J, and the cryostat 

will have to absorb this amount of heat. It is not expected that this will happen very often. In the case of a 

single temporary fault, this will happen only once. In the case of a permanent fault, the amount of energy 

will depend on the number of reclose attempts.

Based on recent data, it is now assumed that a 5 kA, 10 kV superconducting contactor will be 

available by 2025, since superconducting contacts are already available in that current range. High 

temperature superconducting disconnects are presently available with current rating in the 1000s of 

ampere range, and the literature recently reports resistance in the 1 μΩ range (Ref. 32). The most 
NASA/CR—2015-218713 32

promising technique at this date is YBCO with a thin silver plating. Contact resistance on the order of 

1 μΩ has already been reported, with contact force in the range of 10 to 500 N, which is well within range 

of the electromechanical techniques used in conventional power contactors.

Note that “superconducting” is a misnomer for the expected contactor, since the resistance is not zero.

However, it is low enough so that the power losses (0.36 W per contactor at 600 A) are acceptable.

It is expected that the design of the superconducting contactor will resemble that of a conventional 

contactor, except its actuation coil will be superconducting, and its contacts will have very low resistance.

Size and weight of the superconducting contactor will be comparable to that of a conventional contactor.

The actuation coil is expected to have zero steady-state losses. It may or may not have small energy 

losses during opening and closing, but it is not expected that the contactors will be called upon to operate 

very often, so that does not really matter.

Another consideration for this application is contact life (number of switching operations). At present 

there is not much information on how many times superconducting contacts may be operated before 

requiring replacement. The work of Reference 32 indicates that physical degradation of the contacts may 

be an issue. That said, the technology is still in its infancy, so it is quite likely that by 2025 it will advance 

to achieve contact lifetime comparable to that of conventional contacts. Also, in this application, it is not 

expected that there will be any contact erosion due to arcing, since the contacts will be opened only when 

there is no current through them, and will be closed only when the voltage across them is small.

Opening speed is also of interest in this application. In conventional contactors, the usual design 

employs a spring to open the contactor, and an electromagnet to close it. It is possible to increase the 

opening speed by using a spring to close, and an electromagnet to open. In any case, it is expected that 

opening time of around 1 millisecond is possible.

If it is required to open much faster than that, it is possible to use a repulsion disc principle. Because 

the repulsion disc would be superconducting, extremely high forces could be developed, so it would be 

possible to achieve opening times on the order of 100 μsec.
NASA/CR—2015-218713 33

4.0 Design, Operation, and Weights of Hybrid Breakers and Disconnects in 

a TeDP Electric Grid

This section provides details for the design and operation of cryogenic hybrid breakers and 

disconnects for fault isolation and grid reconfiguration in a turboelectric distributed propulsion electric 

grid, using the grid architecture shown in Figure 18.

A viable fault isolation and grid reconfiguration strategy can be based on cryogenic hybrid breakers 

and disconnects with the following features:

• Low steady-state losses

• Rapid fault isolation and grid reconfiguration

• Supports optional reclosing attempts after a fault

During a fault, hybrid breakers provide fault clearing and disconnects provide galvanic isolation and / 

or grid reconfiguration. The strategy can be illustrated with a few fault scenarios.

Figure 18.—Three-Bus Multi-Feeder Architecture with Integrated FCL, Power Conversion, and Circuit Breaker 

Function
NASA/CR—2015-218713 34

The first scenario is a flash-over on one of the feeders, such as on the blue feeder to motor ML-1. In 

this scenario, depending on the details of what triggered the flash-over, it might be possible to clear the 

fault and completely restore the grid simply by tripping and reclosing the DC breaker that supplies power 

from generator GL-1 as well as the DC breaker feeding power to the motor ML-1. The scenario proceeds 

as follows:

1. There is a flashover on the blue feeder to ML-1.

2. DC breakers for GL-1 and ML-1 clear the fault in about 300 μsec. This time includes 100 μsec to 

get the contacts in the breakers open to full gap and begin the interruption process, and 200 μsec

to drive the fault current to approximately zero.

3. After the fault current reaches approximately zero, there will be a small amount of leakage

current, because in the proposed design the breakers do not provide galvanic isolation. In the 

proposed design, the leakage current will have both a DC component and an AC component. The 

DC leakage will be about 2 mA, and is due to the DC leakage through the DC breaker driven by 

the 10 kV DC voltage across the breaker. The AC leakage will be about 20 mA, driven by ripple 

voltage across a capacitor in the breaker included specifically to generate a capacitive current of 

about 20 mA. This creates ideal conditions to clear the fault arc. Because the AC component of 

current is greater than the DC component, there will be current zeros. Because most of the current 

is capacitive, the fault arc will experience “soft switching” in which voltage builds up relatively

slowly after current zero.

4. If the fault clears, it is possible to quickly and completely restore the grid by reclosing the two 

breakers. It is a good idea to reclose the breakers in two stages. During the first stage, the 

electronics in the breakers are turned to the conducting state, but their contacts are not closed.

This enables a very fast “trip-free” re-opening of the breakers if the fault persists. In this scenario, 

it is assumed that the fault is cleared, so the breaker controls proceed to the second stage in which 

the contacts are closed.

5. The second scenario is one in which there is a persistent fault. In that case the scenario starts with 

steps 1 to 3 of the first scenario, but in step 4, the reclose operation results in a trip-free operation 

in which the electronics turn off, and the fault current is interrupted. At that point the DC 

disconnects at either end of the feeder will open to provide galvanic isolation of the fault. Details 

of the interruption of the DC and AC leakage currents will be included in the discussion of the 

details of the operation of the DC disconnects (see Section 4.3).

Another option is to forego the reclose attempt. That is a system control design decision that it might 

be best to defer until actual fault testing can be done. The decision will depend on the probabilities of 

clearing various types of faults.

In the case of a permanent fault, after the fault is removed, the next step is to attempt to restore power 

to motor ML-1. That is done by simply closing the disconnects for the alternate (red) feeder to ML-1, 

with the breaker for ML-1 open to block in-rush current into the electronics for ML-1. There is a remote 

possibility that there is a fault on that feeder, in which case the disconnects will be damaged. In that case, 

breakers will be tripped and reclosed to clear the fault, and ML-1 will be permanently out of service.

Assuming that the disconnects are successfully closed, the next step is to close the breaker for ML-1.

If that is successful, ML-1 is in service. If not, the prudent thing to do would be to trip the ML-1 breaker, 

and take it out of service.
NASA/CR—2015-218713 35

Figure 19.—Proposed Hybrid DC Breaker

4.1 Hybrid DC Breaker

The proposed hybrid DC breaker is shown in Figure 19. Its major components consist of:

• Series inductor, L, which is used to control the rate of rise of fault current, thereby managing the 

amount of current that is flowing when the breaker interrupts the fault, and the amount of energy 

that must be dissipated.

• Cryogenic electromechanical switch to carry load current and momentary fault current. Soft 

switching is achieved by arranging for the primary electronic snubber to be conducting when the 

switch is opened or closed.

• Primary electronic snubber to limit the amount of voltage across the mechanical switch when it 

opens or closes.

• Zinc-oxide surge suppressor, which provides several functions, including:

○ Driving the fault current to zero in the least amount of time after the electromechanical switch 

opens.

○ Limiting the peak transient voltages on the system during fault interruption.

○ Quickly damping post-fault clearing transients to improve the conditions for arc extinction in 

the case of a flash-over.

• Secondary electronic snubber in series with the ZnO surge suppressor, to limit the amount of DC 

leakage current passed by the breaker.

• Shunt capacitor. A small amount of capacitance is connected in parallel with the breaker to 

contribute a small amount of capacitive current to the fault. This will improve the conditions for 

extinction of a flash-over arc.

Operations of the DC breaker include fault interruption, reclosing, and resetting.

Fault interruption proceeds as follows:

First, the fault is detected quickly. Conventional fault detection techniques, including differential 

techniques, can be used to detect the fault in a few tens of microseconds after the fault current exceeds 

maximum normal operating values. Both the primary and secondary electronic snubbers will already be 

turned on; they can be left on continually during normal operation conditions. As soon as the fault is 

detected, opening of the electromechanical contacts is initiated. Contacts will part almost immediately, 

but it will take around 100 μsec to achieve full contact gap and full voltage withstand capability of 20 kV.

Actually, contact parting is a process that takes on the order of 10 to 100 nanoseconds, rather than an 

instantaneous event. During the process, the contact pressure drops from normal design pressure to zero, 

and the contact resistance rises from approximately 1 μΩ to infinity. This creates a brief voltage pulse to 

appear across the contacts that commutates the fault current into primary electronic snubber. As a result 
NASA/CR—2015-218713 36

of the commutation process, some energy will be dissipated in the contacts. It can be shown that the 

amount of energy dissipated is approximately equal to one half of the square of the fault current times the 

inductance of the commutation loop, typically on the order of 100 nH. Contact parting will take place 

almost as soon as the fault takes place, so the current will still be around 1000 A, and the energy 

dissipated by the contacts will be around 0.05 J, which is really a miniscule amount, and should not cause 

any arcing or damage to the contacts.

Current will quickly commutate into the primary electronic snubber as the contacts part. However, at 

that point it is still too soon to switch the primary electronic snubber into a non-conducting state. That is 

because it will take a certain amount of time for the contacts to achieve full voltage withstand. Based on 

experience with repulsion coil actuation, we believe full contact gap can be achieved in approximately 

100 μsec.

Therefore, the controls will deliberately delay 100 μsec after the electromechanical switch starts to 

open before placing the primary electronic snubber in the non-conducting state. Assuming that the value 

of the series inductance is equal to the optimal design value of 1 mH, at that time the fault current will 

have risen from 1000 to 2000 A.

When the primary electronic snubber is placed in the non-conduction state, fault current diverts into 

the shunt snubber capacitor, causing it to rapidly charge to the clamping voltage of the ZnO surge 

suppressor. In order to quickly drive the fault current to zero without creating excessive transient 

voltages, we suggest setting the clamping voltage at 20 kV in a 10 kV system. This will drive the fault 

current to zero in 200 μsec after the electronic snubber goes into the non-conducting state, for a total time 

of 300 μsec to interrupt the fault. At this point in the process the secondary electronic snubber is still 

conducting, so there will be significant (on the order of 2 A) leakage current. In order to drive the leakage 

current still lower, at this point the secondary electronic snubber is placed in the non-conduction state, 

limiting the DC leakage current to a value on the order of 2 mA. By selecting a value of capacitance for 

the shunt snubber capacitor to pass around 20 mA of AC capacitive current, ideal conditions are created 

for the extinction of a flash-over.

The hybrid DC breaker affords the grid designer the option of attempting a fast reclose if the fault is 

cleared. This can be achieved very quickly by simply placing the primary electronic snubber back into the 

conducting state, without reclosing the electromechanical contacts. This enables a reclose without 

incurring a delay for electromechanical reset. If the reclose is successful, the controls can then execute a 

reset operation. If not, the controls can execute a “trip-free” operation by proceeding through the fault 

interruption steps without having to open the electromechanical contacts, since they will already be open.

During the time between reclose and reset, there will be a voltage drop across the breaker on the order of 

1 or 2 V, resulting in a power dissipation of 1 or 2 kW. Therefore, the time delay between reclose and 

reset should be kept as short as possible.

Generally speaking, it is difficult to achieve both fast opening and fast closing of electromechanical 

switches. So, with a 100 μsec opening time, a closing time on the order of 1 msec is a reasonable target.

That will result in a total energy dissipation during the delay to reset of 2 J, which is a reasonable number.

The reset operation itself is trivial. Simply close the electromechanical switch with the primary 

electronic snubber in the conducting state. This will assure that the voltage across the contacts is less than 

about 2 V during contact closing.

We next turn our attention to remaining design questions, starting with the optimal value for the series 

inductor.

The function of the series inductor is to limit the peak fault current and the total amount of energy that 

the ZnO must dissipate. If the value of inductance is too high, the energy due to the initial current 

dominates. If the value of inductance is too low, the energy due to rapidly rising fault current is too high.

There is a value of inductance that minimizes the amount of energy, given by the following equation
NASA/CR—2015-218713 37

 value of load current prior to fault 1000 A

 time delay to open contacts 100 sec

normal operating voltage 10 kV

1mH

0

1

0

0

0 1

= =

∆ = = μ

= =

= ⋅∆ =

I

t

V

I

V t L

(11)

Interestingly enough, with the optimal sized series inductor, the peak fault current is equal to 2 times the 

initial load current.

Also, it can be shown that the total energy dissipated in the ZnO during the fault interruption process 

is given by

 ZnO clamping voltage

energy dissipated in the ZnO

2 0 0 1

0

=

=

⋅ ⋅ ⋅∆ − = ⋅

MAX

MAX

MAX

V

E

V I t

V V

V E

(12)

For the values of design parameters being considered (10 kV system, 1000 A load current, 100 μsec

opening time) and a clamping voltage of 20 kV, the amount of energy dissipated in the ZnO suppressor is

4000 J.

Equations (11) and (12) are based on the approximations that the equivalent circuit model of the 

generators and electronics is an ideal DC voltage source. In reality, the model is more complex than that.

For example, a more detailed model will typically include inductance that will help limit the rate of 

change of fault current. The more complex model will not substantially change the details of the operation 

of the breaker. However, it will reduce the value of inductance needed to limit the rate of rise of current 

and will reduce the amount of energy dissipated in the ZnO suppressors. When it is time to actually 

design and build the grid, it is recommended to perform simulations with detailed models to determine the 

actual values of the hybrid DC breaker circuit parameters.

Also, Equations (11) and (12) were developed for the generator DC breakers. For the motor DC 

breakers, the amount of inductance needed and the amount of energy dissipated in the ZnO suppressors 

will be considerably less than the generator DC breakers. Again, it is recommended to perform 

simulations to determine the actual values needed for the motor breakers.

We have considered alternate methods to the use of the ZnO suppressor to control transient voltages 

during fault interruption, including a shunt snubber capacitor, and have found that the ZnO is the best 

method, for a variety of reasons, including:

• For a given value of peak transient voltage, the ZnO suppressor reduces the fault current to zero 

in less time than any other method. That is because the voltage across the breaker is 

approximately equal to the peak value during the entire time interval during which the fault 

current is going to zero.

• With the use of a ZnO suppressor, residual voltage and current transients are very small 

immediately following the first current zero. This facilitates clearing of flash-over arcing. This is 

in contrast with a capacitive snubber, for example, which produces substantial voltage and current 

transients after the first current zero.

• With the use of a ZnO suppressor, there is very little electrical energy involved in a reclose-reset 

operation. This is in contrast with the use of a shunt snubber capacitor, which will contain a 

substantial amount of energy that must be dissipated in a reclose-reset operation.
NASA/CR—2015-218713 38

We note that in addition to the intentional DC component of the voltage in the grid, there will be a 

ripple component as well, on the order of 500 V at 1000 Hz. This can actually be used to advantage to 

enhance arc extinction in the case of a temporary flash over. There are two reasons for this. If the amount 

of ripple current through the breaker is larger than the DC leakage current, there will be real current zeros, 

which will give the arc a good chance of extinguishing. Also, of the three general types of circuit 

impedance (inductive, resistive, capacitive), capacitive impedance provides “soft-switching” interruption, 

because the instantaneous value of the voltage across the arc immediately after current zero is zero. This 

provides both zero current and zero voltage conditions at the moment of interruption. In order to take 

advantage of this effect, the shunt capacitor should be sized to pass a ripple current of approximately 

10 times the DC leakage current of the breaker. So, for example, with a DC leakage current of 2 mA, a 

ripple current of 20 mA would be reasonable. With 500 V ripple voltage at 1000 Hz, this would require a 

capacitance of about 13 nF, which is quite reasonable.

We also note that the secondary electronic snubber does not need to be rated for full system voltage, 

because of the extreme non-linearity of the ZnO suppressor that it is in series with. For example, through 

careful design, it should be possible to achieve approximately even voltage division between the two 

elements. In order for that to happen, the secondary electronic snubber should not be turned off until the 

voltage across the breaker drops down from 20 to 10 kV.

Finally, we note that measurements of the behavior of metal-oxide varistors have been made at LN2

temperatures (Ref. 33), and no issues have been identified. In fact, performance improves at low 

temperatures, because one of the main considerations in applying metal-oxide varistors is the increase in 

leakage current as temperature rises.

4.2 Loss, Size, and Weight

We now turn our attention to steady-state loss, size, and weight of the hybrid DC breaker. Major 

components include an inductor, cryogenic electromechanical switch, primary electronic snubber, 

secondary electronic snubber, snubber capacitor, and ZnO suppressor.

4.2.1 Weight and Size Estimation for the Series Inductor

The inductor design is sized to limit the rise of a fault current as described in Equation (11). To avoid 

saturation, an air core inductor is typical. A solenoid shape was selected and dimensions were estimated 

based on YBCO tape. The inductance calculation used the approximation below:

l

l d

d L N

1 0.45

1

4

2

2

+

⋅ π ≈ μ (13)

where L is inductance in Henry, N is the number of turns, μ is permeability of air, d is the coil diameter, 

and l is the length of the cylindrical coil. The YBCO current was designed to be 70 percent of critical 

current for the estimated parallel field and coolant temperature. This resulted in an inductor with 

approximate length of 13 cm and a diameter of approximately 17 cm. The weight of the conductor and 

supporting ring was estimated as 6.5 kg.

4.2.2 Cryogenic Electromechanical Switch

The actual contacts in a cryogenic electromechanical switch are a small portion of the total size and 

weight. Most of the size and weight is due to the mechanisms used to achieve high speed operation. The 

design value of the opening time of the cryogenic electromechanical switch of 100 μsec pushes the limits 

of electromechanical technology.

A very good candidate for high speed operation is repulsion coil technology in which a pulse of 

current applied to a pancake coil quickly develops a large force in a copper disk via induced current. This 
NASA/CR—2015-218713 39

approach was used to advantage in a prototype HVDC breaker developed under contract to the U.S. 

Department of Energy in 1978 (Ref. 34). The repulsion disk was used to achieve high speed operation of 

a large valve with an approximate mass of 0.5 kg against a seating force of approximately 2000 N, with 

the valve starting to move in a few microseconds, and significant flow area in 100 μsec. Magnetic 

repulsion was used to high speed opening. A simple spring was used for closing. Tests showed that 

extremely high velocity could easily be imparted to the valve poppet nearly instantaneously. Application 

requirements for the repulsion coil in the HVDC breaker are approximately the same as for the cryogenic 

electromechanical switch, so we expect the size and weight to be approximately the same. Furthermore, 

the physics of the operation of a repulsion coil indicate that its ability to generate large forces improves as 

the temperature is lowered, so we would expect significant improvement in performance at LN2

temperature.

The three largest components of the repulsion coil system were the repulsion coil, the copper disk, 

and the pulse electronics. The electronics were modest in size in 1978; if they were replicated now, they 

would be much smaller, and projection to the time of RTAPS deployment suggests they will be small 

enough to be overshadowed by the coil and disk, which constituted a can shaped cylinder approximately 

8 cm high and 8 cm in diameter, and which weighed approximately 3 kg.

Regarding losses for the repulsion coil system, the only continuous losses are in the charging circuit 

for the pulse generating capacitor, and can be managed to practically vanishing values by using a low 

leakage capacitor.

4.2.3 Primary and Secondary Electronic Snubbers

4.2.3.1 Requirements

Requirements were explained in the previous sections. It is assumed that the grid can be designed so 

that the electronic snubbers are unidirectional. If that is not that case, then we will have to double up, and 

use two sets of snubbers to handle the bidirectional case.

Due to the fact that snubber 1 in the breaker needs to conduct a peak current of 2 kA, exhibit a 

voltage drop of less than approximately 3 V at 1 kA, and be able to block 20 kV, makes it the more 

difficult to design than snubber 2.

4.2.3.2 State-of-the-Art

Currently available IGBT silicon devices have a maximum blocking voltage capability around 6.5 kV 

and a peak current capability of about 1.5 kA, and exhibit a voltage drop of about 3.5 to 4.5 V at 1 kA at 

room temperature. This makes them not a good option for this application. Example parts include 

Mitsubishi/Powerex CM750HG-130R and Infineon FZ750R65KE3.

Cryocooling these existing devices will de-rate the maximum blocking voltage to about 70 percent of 

the room temperature capability and also decrease the conduction voltage drop. In order to block 20 kV, 

5 IGBT in series will be required, which will require a voltage drop per device equal to 0.6 V at 1000 A in 

order to meet the specifications. Given the published data on these two switches, cryocooling will 

decrease the voltage drop per device from about 3.5 to 4.5 V at 1 kA at room temperature to about 1.5 to 

2.5 V at 1 kA at cryogenic temperature. This is not adequate to meet the 0.6 V required.

Thyristor technology, on the other hand, if cryocooled, can provide the desired performance in terms 

of blocking voltage capability and forward voltage drop at 1 kA. State-of-the-art thyristors can block 

voltages as high as 8 kV and conduct currents as high as 2.3 kA (Infineon T2563 NH and T600N).

If cryocooled, their voltage blocking capability can still be considered to be about 70 percent of the 

rated voltage, making it capable of withstanding 5.6 kV. Therefore, only four devices will be needed.

Three volts distributed over four devices connected in series translates to a 0.75 V drop per device at 1 kA 

when cryocooled. Given the performance of the parts at room temperature (drop of 2.75 V at 5 kA), the 

required performance of 0.75 V cryocooled becomes very feasible. Unfortunately the thyristor cannot be 

easily turned off, which is a must for this application.
NASA/CR—2015-218713 40

4.2.3.3 Future Developments

It is clear that high level of doping, and a proper doping profile, lead to a part with large voltage 

withstanding capability as well as very low conduction voltage drop. This has been demonstrated for the 

thyristor already. Very large levels of doping have been avoided for IGBTs structure due to fact that it 

will lead to slow turn-off transitions. However, for this application, turn-off requirements are not that 

demanding. Typical IGBT turn off times are on the order of 0.1 μsec, while all that is needed for this 

application is turn-off less than about 20 μsec, so there is plenty of margin. Furthermore, cryocooling will 

increase the turn-off speed and decrease the conduction voltage drop making it possible to extrapolate 

that, in the near future, for cryogenic applications, an IGBT with such doping concentrations and doping 

profile will exist and will be comparable in size to an existing thyristor.

4.2.3.4 Weight and Size Estimation for Electronic Snubber 1 (20 kV, 3 V at 1 kA, max 2 kA)

Using the thyristor size and mass we can extrapolate that a stack of four devices will be a cylinder about 

200 mm long, and have a diameter of 175 mm approximately, and an estimated weight of about 16 kg.

4.2.3.5 Weight and Size Estimation for Electronic Snubber 2 (5 kV, max 2 kA)

This snubber will require only one element; therefore, we can extrapolate that a single device will be 

a cylinder about 50 mm long, and have a diameter of 175 mm approximately, and an estimated weight of 

about 4 kg.

4.2.4 Weight and Size Estimation for Snubber Capacitor

The value of the snubber capacitor will be on the order of 10s of nanofarads, which is quite small, 

even with a 10 kV DC and 20 kV peak rating. Order of magnitude size and weight is less than that of a 

hockey puck.

4.2.5 Weight and Size Estimation for Zinc-Oxide Suppressor

Since the ZnO suppressor will be operating in a contamination-free environment, we assume that it 

can be comprised primarily of ZnO material, without the need for the ceramic insulators that are included 

in the design of station surge arrestors for protecting the ZnO from contamination.

The key design parameters for the ZnO suppressor are peak electric field, peak current density, and 

total energy dissipated. From Reference 33 we conclude that there will not be any issues with operation at 

LN2 temperatures. From Figure 3 of Reference 35, we can see that it is possible for a ZnO suppressor to 

support an electric field of 2000 V/cm, and a current density of 10 A/(cm*cm). We will come back to 

energy dissipation later.

For a nominal load rating of 1000 A and 10,000 V, and switch opening time of 100 μsec, the previous 

analysis indicates a peak voltage across the ZnO of 20,000 V, and a peak current of 2,000 A. If we 

combine these values with the supported electric field and current density, we find the required length is 

10 cm, and the required area is 200 cm2

, for a total volume of 2000 cm3. The density of ZnO is 

approximately 5.6 grams/cm3

. So, based on peak current and voltage considerations, the weight of the 

ZnO is 11.2 kg.

We now go back and check the energy dissipation. The analysis in the previous sections indicates that 

there will be about 4000 J dissipated in the ZnO when the DC breaker operates. This comes out to be 

2 J/cm3

, which is very low for ZnO. To put that in perspective, typical energy dissipation capabilities of 

station arrestors are on the order of 2.6 kJ/kV, or 26,000 J at 10 kV. So there is plenty of margin with 

respect to energy dissipation; in this application all that we need to worry about is peak electric field and 

peak current density.
NASA/CR—2015-218713 41

4.2.5.1 Hybrid DC Breaker Preliminary Sizing Model

Table 7 provides the basis for a sizing scalar. Major component masses were estimated for a design 

point, which then allows calculation of a kilowatts/kilograms scalar of 187 kW/kg if no packaging or 

containment is considered. As will be described in later sections, if high pressure cryogen is assumed, this 

significantly reduced the power density to ~140 kW/kg, as shown in the table. The containment mass of 

18.5 kg assumes an aluminum capsule designed for 650 psi cryogen.

The loss model assumes that the experimental values of ~1 μΩ can be achieved at current levels, 

which would span the range of the study voltages.

4.3 DC Disconnect

The circuit for the DC disconnect is shown in Figure 20. It has a relatively simple design because the 

electrical behavior of the DC breaker creates nearly ideal soft-switching conditions for the opening of the 

disconnects. The residual current through the breaker has a DC component of about 2 mA, and an AC 

component of about 20 mA through the breaker’s shunt snubber capacitor. The disconnect consists of 

three elements: a cryogenic contactor, a snubber capacitor, and a snubber resistor. The contactor offers 

low resistance (approximately 1 μΩ) carrying of normal load current. The snubber capacitor assists 

interruption of residual DC current, and the snubber resistor prevents a spike of current through the 

contactor when it closes. The two functions of the DC disconnect are to open to isolate a fault and to close 

to reconfigure the grid subsequent to fault isolation.

TABLE 7.—POINT DESIGN SUMMARY

Design Point Sizing

Rated Current ........................................................ 1,000 A

Rated Voltage...................................................... 10,000 V

Power ............................................................... 10,000 kW

Major Components

Electronics snubbers..................................................20 kg

ZnO ...........................................................................10 kg

Switch .........................................................................3 kg

Inductor (below)................................................... 11.42 kg

Subtotal ................................................................ 44.42 kg

Bracket and wiring adder ............................................20%

Bracket and wiring adder .......................................8.88 kg

Minimum Component Volume.............................0.012 m3

Containment Volume ...........................................0.042 m3

Containment mass................................................ 18.53 kg

Total Mass.................................................................71.84

Figure 20.—DC Disconnect
NASA/CR—2015-218713 42

In the case of a permanent fault, the DC breaker reduces the current from fault levels of thousands of 

A to DC and AC leakage currents of around 2 and 20 mA, respectively. If it is determined that the fault 

has not cleared, the appropriate DC disconnects are opened to complete the isolation process. They 

simply open. The leakage current will quickly commutate into the RC snubber. The amount of energy 

dissipated by the contacts during the very brief time interval that the contact resistance rises to infinity is 

determined mainly by the stray inductance of the commutation loop. An order of magnitude computation 

based on 100 nH and 20 mA indicates the amount of energy dissipated is on the order of 20 pJ, which is 

miniscule. So, the opening duty is very mild. The general shape of the switching waveforms is shown in 

Figure 21. At the time of contact opening, the current through the disconnect has been reduced to 2 mA

DC resistive leakage, and 20 mA AC ripple at 1000 Hz.

No effort is made to synchronize the opening of the disconnect with the current zeros; it is not 

necessary to do so. At the instant the contacts part, whatever current that is flowing through them is 

quickly transferred to the RC snubber, so on the time scale of Figure 21, the current through the contacts 

makes a step change to zero. At that point the current flows through the RC snubber. There is a very small 

step in voltage, on the order of 2 V, generated by the step change of current through the snubber 

resistance. The major portions of the voltage are from the DC and AC components of the leakage current. 

The instantaneous contributions of these two sources at the instant of contact parting are both zero, 

because the voltages on the various capacitors in the circuit cannot change instantaneously. The AC 

component of the recovery voltage results from a voltage divider action between the capacitors, with an 

instantaneous value at contact parting of zero. The DC component of the leakage current produces an 

exponential decay in current as it charges the disconnect snubber capacitor, so there is a component of 

voltage across the contacts that rises exponentially to 10 kV. There is also a small contribution to the 

recovery voltage as a result of the interaction between the disconnect snubber capacitor and the breaker 

inductance, but it is small because the product of the residual leakage current times the equivalent surge 

impedance (square root of L/C) of the snubber capacitor and breaker inductance is small. Therefore, the 

major contribution to the recovery voltage imposed on the disconnect after its contacts separate results 

from charging the snubber capacitor via the breaker leakage current.

In order to determine how large C must be, we must consider the DC leakage current, and how 

quickly we can allow the 10 kV to appear across the contacts after they part. It should be possible to 

achieve full contact gap in 100 μsec, so the time constant formed by C and the equivalent leakage 

resistance should be set appropriately. A time constant of around 1 msec should be more than adequate. 

With a DC leakage current of 2 mA at 10 kV, the equivalent leakage resistance is 5 MΩ. To achieve a 

1 msec time constant, C should be around 200 pF, which is a very small value.

Figure 21.—Current and Voltage During the Opening of Disconnect
NASA/CR—2015-218713 43

In order to determine R, we must consider how much voltage we can allow across the switch as it 

opens, and how much current we can allow as it closes. With a target of 2 V during opening, with 20 mA, 

R should be approximately 100 Ω. During closing, with the snubber capacitor charged to 10 kV, there 

will be a current pulse with 100 A peak and time constant of 0.02 μsec, which does not present any issues.

4.3.1 DC Contactor Preliminary Sizing Model

Table 8 provides the basis for a sizing scalar. Major component masses were estimated for a design 

point, which then allows calculation of a kW/kg scalar of 1540 kW/kg. It is notable that containment and 

packaging mass are estimated to be quite low due to the small volume of active components.

The loss model assumes that the experimental values of ~1 μΩ can be achieved at current levels, 

which would span the range of the study voltages.

4.4 Hybrid AC Breaker

There are two general approaches on implementing a hybrid AC breaker for use in the propulsion 

electric grid, depending on whether or not we are looking for a fault-current-limiting breaker, or if all we 

need is interruption of the fault current at the first naturally occurring current zero. If current-limiting is 

required, the thing to do is to use three of the hybrid DC breakers shown in Figure 19 for three phase 

current-limiting AC service as well, with some minor changes. For AC service, the logic for sizing the 

components in Figure 19 changes. It is recommended to do some simulations when it comes to 

implementation time to determine appropriate sizes of the components. In the case of AC service, residual 

leakage current through the breaker after it operates is on the order of 20 mA AC, so there is an excellent 

chance that a temporary fault will clear. If it does clear, the breaker can execute a fast reclose / reset 

operation as described for the DC breaker. If it does not clear, then the backup strategy is to turn off the 

generator and turn it back on. If that fails, then the generator will have to be taken out of service.

If current-limiting is not required, then the hybrid AC breaker can be implemented using thyristors.

One pole is shown in Figure 22. There is a cryogenic contactor to carry load current during normal 

conditions, two anti-parallel thyristors to provide soft switching of the contactor, and possibly a snubber 

capacitor to limit the rate of rise of recovery voltage across the thyristors. In this case, a ZnO suppressor 

is not needed, because operation of the breaker does not trap any inductive energy.

TABLE 8.—CONTACTOR DESIGN POINT SIZING

Rated Current........................................................1,000 A

Rated Voltage .....................................................10,000 V

Power...............................................................10,000 kW

Major Components

Switch ........................................................................ 3 kg

Snubber RC............................................................1.00 kg

Subtotal..................................................................4.00 kg

Bracket and wiring adder........................................... 20%

Bracket and wiring adder.......................................0.80 kg

Volume .................................................................... 0.004

Containment mass..................................................1.69 kg

Total Mass ................................................................. 6.49
NASA/CR—2015-218713 44

Figure 22.—Proposed Hybrid AC Breaker

Figure 23.—Switching Devices in Architecture 1

Under normal conditions, the thyristors are gated on, just to be always ready. In the case of a fault, the 

contactor is opened. In this case, speed of the contactor is not as important as for the DC case, but we will 

assume the use of the same contactor as for the DC case, since it will be available, and that the time 

between contact parting and fully open contacts is 100 μsec. So, 100 μsec after contact parting, gating of 

the thyristors is turned off, and the fault is cleared at the first natural current zero.

The thyristors are sized for full system voltage, and for 1⁄2 cycle of full fault current.

4.5 Breaker and Disconnect Locations

Figure 23 shows Architecture 1 with eight switching devices by number. The types of those switches 

are described below.

• Switch 2, 6, and 8 are hybrid DC breakers

• Switch 3, 4, 5 are DC disconnects

• Switch 1 and 7 are hybrid AC breakers

All of the three types are addressed in detail in the previous sections.
NASA/CR—2015-218713 45

4.6 Hybrid DC Breaker Operation in the System

This section provides more detail for two aspects of the operation of cryogenic hybrid breakers during 

fault interruption in a turboelectric distributed propulsion electric grid that were observed during 

simulations:

1. For a fault at the terminals of a motor, the fault current flows through two hybrid breakers. As a 

result, there are two inductors instead of one inductor in series during the rise and interruption of 

fault current. Consequently, compared to a fault at a generator bus, the fault current rises and falls 

more slowly and fault clearing time and dissipated energy increases a little bit.

2. After the fault current falls essentially to zero, the incremental resistance of the ZnO snubber is 

very high. As a result, there is very little damping for the voltage oscillations produced by the 

interaction of the snubber capacitor and the series inductor, and a ringing voltage is produced 

after the interruption is complete.

4.6.1 Motor Faults

The relevant circuit for a motor fault is shown in Figure 24. There are two inductors in series with the 

fault. This will lower both the rate of rise of current prior to breaker operation and the rate of fall after the 

breaker opens. This will have the following consequences:

1. The amount of current flowing when the breaker opens will not be as large as for a bus fault.

2. The amount of time required to drive the current to zero will be slightly larger than for a bus fault.

3. The amount of energy dissipated by the ZnO snubber will be slightly greater than for a bus fault.

It is useful to examine four relevant cases:

1. Bus fault at full voltage.

2. Bus fault at reduced voltage.

3. Motor fault at full voltage.

4. Motor fault at reduced voltage.

For the purposes of this discussion, full voltage is 10 kV and reduced voltage is 8 kV. The ZnO is 

designed to clamp at 20 kV. The series inductor is 1 mH. Nominal load current is 1 kA. The time between 

fault detection and breaker operation is 100 μsec.

Figure 24.—Circuit Condition for Motor Fault

Motor
NASA/CR—2015-218713 46

For a bus fault at full voltage, 10 kV, the current starts from 1 kA and rises to 2 kA, 100 μsec after the 

fault. The breaker operates and the ZnO clamps at 20 kV. This applies 10 kV across the inductor, 

producing a rate of change of current of negative 10 A/μsec. It takes an additional 200 μsec for the 

current to fall to zero. During that time, 4000 J are dissipated in the ZnO. In summary:

• System voltage: 10 kV

• Fault location: bus

• Total time to operate: 300 μsec

• Peak current: 2 kA

• Energy dissipated in ZnO: 4000 J

For a bus fault at reduced voltage, 8 kV, the current starts from 1 kA and rises to 1.8 kA, 100 μsec

after the fault. The breaker operates and the ZnO clamps at 20 kV. This applies 12 kV across the inductor, 

producing a rate of change of current of negative 12 A/μsec. It takes an additional 150 μsec for the 

current to fall to zero. During that time, 2700 J are dissipated in the ZnO. In summary:

• System voltage: 8 kV

• Fault location: bus

• Total time to operate: 250 μsec

• Peak current: 1.8 kA

• Energy dissipated in ZnO: 2700 J

For a fault at the motor, there are two inductors in series. With full system voltage, 10 kV, for a fault 

at the motor terminals, the fault current through the breaker rises at a rate of 5 A/μsec to a peak current of 

1.5 kA. The breaker operates and the ZnO clamps at 20 kV. This applies 10 kV across the pair of 

inductors, producing a rate of change of current of negative 5 A/μsec. It takes an additional 300 μsec for 

the current to fall to zero. During that time, 4500 J are dissipated in the ZnO. The two inductors form a 

voltage divider that places 15 kV on the generator buss. In summary:

• System voltage: 10 kV

• Peak bus voltage: 15 kV

• Fault location: motor

• Total time to operate: 400 μsec

• Peak current: 1.5 kA

• Energy dissipated in ZnO: 4500 J

Finally, with a system voltage of 8 kV, for a fault at the motor terminals, the fault current through the 

breaker rises at a rate of 4 A/μsec to a peak current of 1.4 kA. The breaker operates and the ZnO clamps 

at 20 kV. This applies 12 kV across the pair of inductors, producing a rate of change of current of 

negative 6 A/μsec. It takes an additional 233 μsec for the current to fall to zero. During that time, 3266 J

are dissipated in the ZnO and 13 kV appears at the generator bus. In summary:

• System voltage: 8 kV

• Peak bus voltage: 13 kV

• Fault location: motor

• Total time to operate: 333 μsec

• Peak current: 1.4 kA

• Energy dissipated in ZnO: 3266 J
NASA/CR—2015-218713 47

Comparing the full voltage bus fault with the full voltage motor fault we see that the energy 

dissipation is increased slightly from 4000 to 4500 J, and the total operating time is increased from 

300 μsec to 400 μsec.

4.6.2 Ring Down

During simulations, a ringing transient voltage was observed after interruption. Figure 25 is a typical 

example. The top figure is the voltage across the ZnO and the lower figure is the current through the ZnO 

during ZnO operation. The mechanism producing the transient is the interaction between the snubber 

capacitor and the series inductor after the ZnO stops conducting. The question arose whether or not the 

transient is a real effect, or perhaps it was an artifact of the simulation. Analysis shows that for the circuit 

used, some ringing is expected, but it is not likely to be as pronounced as shown. For the actual circuit, its 

amplitude will depend on subtleties that would be difficult to model in a simulation, such as the values of 

stray capacitance and stray inductance in the circuit, and the high frequency losses of the series inductor.

Although there are no obvious problems that the ringing might engender, intuitively it would seem to 

be a good idea to prevent them if possible. There is a relatively easy way to do so, by increasing the high

frequency losses of the series inductor. This can be done by connecting a resistor in parallel with the 

inductor. The breaker circuits before and after the addition of the resistor are shown in Figure 26 and 

Figure 27. The much cleaner post-interruption voltage waveform is shown in Figure 28.

Figure 25.—Typical Voltage Ringing after ZnO Suppresses Overvoltage

Figure 26.—Original Hybrid DC Breaker Circuit
NASA/CR—2015-218713 48

Figure 27.—DC Circuit Breaker with the Additional Resistance

Figure 28.—Effective Voltage Waveform After Adding Additional Ramping Resistance

The value of the resistance is not critical; there are a broad range of values that will work. It should 

not be so low as to significantly add to the fault current, but it should be low enough to provide damping.

An order of magnitude computation indicates it should be approximately equal to the square root of the 

ratio of the series inductance over the snubber capacitance, which is about 30,000 Ω. However, since it is 

likely that stray capacitance is likely to be several picofarads, we may wish to raise the value of the 

snubber capacitance a bit, which would lower the required value of resistance. It could be dropped down 

quite a bit from 30,000 Ω without substantially raising the fault current. For example, a value of 1,000 Ω

would provide plenty of damping, with a contribution to the fault current of only 10 A.

Furthermore, the steady-state rating of the resistor is zero, since it normally does not carry any 

current. What does matter is the amount of energy it will absorb. For a 1,000 Ω resistor, the dissipation 

during fault interruption is only 20 J.

Though potentially a minor effect, the desire to add this resistor was identified through system

simulation discussed in the following sections.
NASA/CR—2015-218713 49

5.0 Transmission Cable Considerations

Initially superconducting transmission cable mass estimates assumed only the superconducting core 

mass per unit length. This section addresses some assumptions to allow estimation of the transfer line, 

which refers to the line construction to carry cryogen and provide insulation through a vacuum jacket.

Though the cooling system was not part of the study scope, some assumptions and calculations were 

required to make an estimate. As the reader will see in this and following sections, the transfer line mass, 

at the existing technology level, adds a significant mass to the transmission system mass estimate.

Based on the early analysis and literature on cryogenic semiconductors, a two operating cooling fluid 

system was assumed for the purpose of progressing system weight estimates related to the transmission 

lines. The cooling system is summarized below.

• Power Conversion Operating at ~150 K: Selected to minimize system losses (electrical and 

cooling).

• Electric Machines Operating at ~30 to ~60 K: Selected based on the assumption of machine 

incorporating state-of-the-art HTS into machines that would achieve TRL 4 to 6 in the 2025 

timeframe, given that these HTS materials require cooler temperatures for moderate fields. It is 

recognized and acknowledged that ventilation of the same electric equipment enclosure with 

electronics and machinery cooled by different cryogens at different thermodynamic state will 

introduce heat transfer challenge requiring a novel approach. Our assumption is that this 

challenge is not a show-stopper, and will avail itself to a practical engineering solution.

If these assumptions change and technologies advance, it is still anticipated that the models presented 

will provide an approach to updating estimates. For example, a single coolant at another temperature 

could be selected and the system resized.

Lastly, in order to estimate the weight and thermal (heat leak and pumping pressure drop) losses of 

the transmission system, the cooling approach and fluids were required. Use of a supercritical fluid was 

assumed, referencing earlier DOE-funded, Fusion Energy studies (Refs. 36 and 37) of cryogenic 

convection heat transfer along elongated flow paths with distributed heat input For cryogen flow in 

transfer piping, loop pressurization to the supercritical state is attractive compared to saturated or 

subcritical states because it avoids two-phase flow transition events and associated hydrodynamic 

instabilities that can be prevalent in other thermodynamic regimes. Forced convection heat transfer 

coefficients on the order of 1 kW/m2

-K are achievable with supercritical flows.

Furthermore, by tuning flow rate and path inlet conditions, supercritical flows can be engineered to 

resist temperature rise (or even cool) as they absorb heat from external sources (Refs. 38 and 39).

Supercritical N2 and Ne were selected for the power electronics and superconducting loops respectively.

If need be, the superconducting loop model / sizing can be updated for lower temperature operation with 

hydrogen to enable machine sizing at increased power density.

Two submodels were developed: one for the supercritical N2 flow loop, cooling both generator and 

motor power conditioning electronics, the other for the Ne-cooled superconducting transmission cable 

and machinery windings. Both submodels include a 1-D flow network and associated heat transfer 

calculus to estimate cryogen pressure drop and temperature change from inlet to outlet, based on user 

inputs as well as parameter value exchange with the master system sizing model. The N2 submodel is 

informed by vacuum-insulated, corrugated transfer tube technology (commercially available, with design 

information embedded in the model). The other submodel is similar, but substitutes Ne for N2 and adds 

HTSC transmission line technology (commercially available, with design information embedded in the 

model). In both submodels, the commercial component information includes key design geometry, bend 

radii limits, ampacity ratings, heat leak, and weight values as look-up tables or curve fits. Accordingly, 

the TeDP sizing studies are limited by interpolation within the present range of embedded values.
NASA/CR—2015-218713 50

Cryogen supply conditions are also constrained based on the currently-hypothesized temperature and 

pressure ranges. If needed, that range can be shifted or expanded by updating the submodels.

Figure 29 and Figure 30 show the N2 and neon cooling system layouts used for estimation of coolant 

flow, transmission line sizes, heat leak, pressure drop, and pumping power. As can be seen, central manifold 

and breaker boxes are assumed; this will likely not be the case in practice when designing for redundancy, 

but it provides a reasonable basis for estimating transfer line lengths and size. These configurations are used 

only to estimate cable and transfer line estimates, but no other equipment shown on the diagram. Additional 

assumptions used with these structures for system estimates are listed below:

• N3-X fuselage dimensions

• 213 ft wingspan:

○ Power plants / generators located at wing tips.

○ Propulsion fans / motors proximal to airframe center.

• Pristine DC power output from generators. Zero harmonic content.

• Circuit breakers cooled by Ne.

• Ohmic losses of HTS splices are negligible.

• Landing shock, aircraft maneuver, and wing flexure do not quench HTS bus.

• Electrical intertie between neon-cooled and N2-cooled circuits established by resistive current 

leads that are conduction-cooled (passive heat-sinking; no cryocooler-anchored thermal intercept 

station).

Figure 29.—Nitrogen Cooling System

Figure 30.—Neon Cooling System

Nitrogen

Refrigeration Plant

Valve / Manifold

Box

Generator

Power 

Electronics

Generator

Power 

Electronics

Generator

Power 

Electronics

Generator

Power 

Electronics

Fan Motor Power Electronics

Nitrogen Supply Nitrogen Return

Nitrogen Supply Nitrogen Supply

Nitrogen Return Nitrogen Return

Neon

Refrigeration Plant

Valve / Manifold

Box

Generator

HTS Circuits

Generator

HTS Circuits

Generator

HTS Circuits

Generator

HTS Circuits

Fan Motor HTS Circuits

Breaker Box

...32 +/- Line Pairs

64 Triax

...16 Lines

2 +/- Line Pairs

4 Triax

2 +/- Line Pairs

4 Triax

Neon Supply Neon Supply

Neon Supply Neon Return
NASA/CR—2015-218713 51

The following assumptions resulted in a significant mass increase compared to the HTSC core only 

estimates made earlier in the study. Length estimates increased significantly due to bend radius 

assumptions. The mass estimates shown in Table 9 assume an 8 kV bus voltage. Mass per unit length for 

both the transfer line and superconductor core are included and are based on the power requirements of 

the cable lengths estimated and the masses listed in Section 6.6.

TABLE 9.—TRANSMISSION AND 

DISTRIBUTION MASS ESTIMATE

Ne System

HTSC................................................................... 1070 kg

Transfer Line...........................................................3477 kg

Ne Cryogen.............................................................210 kg

N2 System

Transfer Line ............................................................432 kg

N2 Cryogen ..............................................................81 kg

Total.............................................................................5270 kg
NASA/CR—2015-218713 52

6.0 Component Sensitivities to Voltage

6.1 Sensitivity of Components to System Bus Voltage

The first order mass and efficiency sensitivity of components to bus voltage is simply summarized in 

Table 10. This table assumes a given power level and a voltage range from approximately 1 to 

approximately 10 kV.

Electric machines have already been shown to scale with power; the remaining three component 

classes are discussed below. At the end of this section, a summary is provided of mass efficiency scalars 

that can be used for study.

6.2 Sensitivity Analysis of Weight and Efficiency for Transmission Lines

The transmission lines are weight-sensitive to a voltage change at fixed power example. A decrease in 

voltage results in an increased current requiring more current-carrying superconductor. It should be noted 

that a significant portion of mass associated with state-of-the-art superconducting transmission lines is 

associated with cooling and insulation of the line. Estimates including the construction for cooling and 

insulation become more complicated as it involves feeders of various power levels, cooling flow 

requirements, and step changes in the overall size of tubes. These were provided as part of a study sizing 

tool spreadsheet, but a plot, Figure 31, can be used as a generalization.

In Figure 31, transmission line mass is shown to decrease with increasing voltage. In addition to the 

HTSC mass, the transfer line that carries the cryogen and provides the vacuum jacket must also be 

included. Across the range of voltages considered this was estimated as 4 kg/m. Using these numbers and 

an HTSC line length of 700 m, the mass as a function of voltage can be estimated. This will be similar to 

the spreadsheet tool, but not identical due to the tool’s additional detail in transfer line mass estimates and 

lines of different ratings supplying to different components (motors and generators).

The reader will note that the majority of the mass is the transfer line, not the HTSC core.

6.3 Sensitivity Analysis of Weight and Efficiency for RTAPS Hybrid Breakers / 

Disconnect Switches

To a first approximation, the size, weight, and losses of RTAPS hybrid breakers and disconnects 

depends only on rated power (rated voltage time rated current), with no dependence on system voltage, 

over a modest range of system voltage, such as from 5 to 50 kV.

At a high level, this can be seen as a result of the fact that it is possible to design a building block 

breaker or disconnect module of a certain voltage and current rating, and then connect the modules in 

series / parallel to achieve a higher voltage and current rating. Suppose that the system voltage is M times 

the voltage rating of the building block, and system current is N times the current rating of the building 

block. Then, M*N building blocks will be required, M in series, N in parallel. The overall size, weight, 

and losses will be M*N times the size, weight, and losses of a single building block.

TABLE 10.—FIRST ORDER SENSITIVITY FOR

VOLTAGE RANGE OF ~1 TO ~10 kV BUS VOLTAGE

Component Weight sensitivity Efficiency sensitivity

Electric machines No No

Power conversion Yes Yes

Protection Breakers and Contactors No Yes (small impact)

Transmission lines Yes No
NASA/CR—2015-218713 53

Figure 31.—HTSC Core and Total Transmission Line Mass (700 m) vs. DC Link Voltage

It is useful to explore the design of two of the major components of the hybrid breaker: the ZnO surge 

suppressor, and the electronic snubbers.

For the ZnO, the key design constraints are the maximum allowable electric field and current density 

in the material. Consider a ZnO cylinder. The required area of the cylinder is equal to the peak current 

divided by the maximum allowable current density. The required height of the cylinder is equal to the 

peak voltage divided by the maximum allowable electric field. Therefore, the size and weight of the 

cylinder is equal to the peak current times peak voltage divided by the product of maximum allowable 

current density times maximum allowable electric field. In other words, the size of the ZnO is 

proportional to the rated power of the grid being protected.

A similar argument, based on electric fields and current density, holds true for the electronic 

snubbers.

For the contacts themselves, several identical contacts can be placed in series and parallel to achieve a 

given voltage and current rating. It can be shown that the total losses will be proportional to the number of 

contacts needed.

The only components of the breakers and disconnects that may depend on system current rating are 

the superconducting connections between other components. Since their sizing depends mainly on 

current, as the system voltage rating goes up, for a given power, the current rating will go down, so the 

size of the superconducting connections should go down. However, the size and weight of the 

connections is a small fraction of the total for the breakers and disconnects, and can be ignored in a first 

approximation.

6.4 Sensitivity Analysis of Weight and Efficiency for Power Converters

The purpose of this effort is to understand the impact of the DC link voltage to the weight and 

efficiency of the power converter under cryogenic environment. Three major components of the power 

converter are considered: power stack, passive filter, and cooling system. Three-level Neutral-PointClamped (NPC) converter topology (as shown in Figure 32) is selected for the study, since it is widely 

used in the power industry for medium voltage power conversion application.

0 2000 4000 6000 8000 10000 12000 14000

0

1000

2000

3000

4000

5000

6000

Vdc (V)

Mass (kg)

HTSC Transmission Line Total

HTSC core only
NASA/CR—2015-218713 54

Figure 32.—Three-Level NPC Converter

6.4.1 Power Stack

The power stack consists of the semiconductor devices, the auxiliary circuits, and the bus bars. As 

discussed in the previous report, Si-based IGBTs are preferred in cryogenic environment due to its 

improved loss performance. The output characteristics curve of the IGBT will be very steep at low 

temperature, which means the forward voltage drop is not sensitive to the current flowing through the 

device. In this study, the forward voltage drop of the IGBT is assumed to be 1 V per IGBT, regardless of 

the voltage rating of the device. The switching loss of IGBT will also be reduced significantly at

cryogenic temperature, mainly due to the reduced tailing time (Ref. 40) and the improved diode reverse 

recovery performance (Ref. 41). The turn-on loss and turn-off loss at cryogenic condition are assumed to 

be reduced to 50 and 20 percent, respectively compared to operation at room temperature. Three 

commercial IGBTs (FZ750R65KE3, FZ1200R45KL3, FZ1200R33KL2C) are used as the benchmark to 

extrapolate how the switching loss and the power density vary with the device rated voltage. Although 

mature and widely used in the high power industry, the technology of Si IGBT is still evolving (Ref. 42).

This results in a projection of 20 percent improvement for the switching loss is assumed in the study.

For device voltage rating selection, a safety factor of 0.6 is applied. For example, if the voltage across 

the device is 2 kV when the device is turned off, then a 3.3 kV device will be selected. The maximum 

single device rating is assumed to be 10 kV. If the required blocking voltage is higher than 12 kV, two 

devices will be put in series as one switch unit.

In this study, the switching frequency of the IGBT is considered to be a function of the device rated 

voltage. In general, the device with higher voltage rating will have lower switching speed and higher 

switching loss, and, therefore, the maximum feasible switching frequency will be lower. Considering the 

common practice, we choose 1.8 kHz for the 3.3 kV device and 600 Hz for the 10 kV device. Then the 

switching frequency for any other voltage rating can be extrapolated. The fundamental frequency is 

assumed to be 60 Hz in the study. Since the ratio of the switching frequency over the fundamental 

frequency is not high, the switching frequency is rounded to be the integral multiple of the fundamental 

frequency in order to avoid subharmonics. Therefore, the variation of the switching frequency is not 

continuous.

6.4.2 Passive Filter

The design of passive filters involves many different aspects of the system, including system stability, 

standards applied to the system, hardware layout, control and modulation scheme, etc. Some of the 

aspects require very detailed converter design, e.g., the EMI consideration, since it is related to not only 

the converter voltage / current / control setting, but also the parasitics inside the converter and the ground 

connection. In this study, we choose Total Harmonic Distortion (THD) and energy storage as the criteria 

for sizing the passive filter. This approach avoids the excessive details of the converter hardware design, 

and it represents a good lump-sum estimation of the passive components for the sensitivity study.

Vdc/2

Vdc/2
NASA/CR—2015-218713 55

Spectrum analysis is performed to extract the harmonic voltage content at the output of the 3-level NPC 

converter, with which the required inductance to meet certain THD requirement can be calculated. Five 

percent THD is assumed in the study. For the DC link capacitor, its main function is to maintain the 

stability of the bus and minimize the voltage ripple. The energy storage requirement is applied for the 

capacitance selection (Ref. 43). In this case, we assume that the DC link voltage will not drop below 90

percent of the rated value even if the input power was interrupt for three switching cycles.

The sizing of the inductor follows the same approach as the one used to size the inductor of the hybrid 

circuit breaker. It is an air core design. The sizing of the capacitor is based on energy storage. A film 

capacitor is selected due to its good energy density and reliability. The state-of-the-art energy density of 

the film capacitor is 0.4~1 kJ/kg. A 5 kJ/kg is used in the sensitivity study as future projection (Ref. 44).

6.4.3 Cooling System

The power stacks (including the DC link capacitors) of the converter are installed in a cryogenic tank 

or vessel in order to maintain the low operating temperature for the semiconductor device and minimize 

interconnection between difference interfaces. All the electronics components are assumed immersed in 

the cryogen. The weight and volume of the coolant and the tank depend on the volume of the power 

stacks and the package fill factor. The fill factor represents the fraction of volume that is occupied by the 

power stacks. The fill factor of the present medium voltage converter is below 0.2. For this aerospace 

case, a projection of 0.4 is used.

The approach to size the tank is the same as the one used for the containment of the hybrid DC circuit 

breaker.

6.4.4 Conclusion

Figure 33, Figure 34, and Figure 35 show results for the voltage sensitivity study. The saw tooth 

shape ripple is due to the quantization effect of the switching frequency, as described in the power stack 

section.

From 2 to 12 kV, only one IGBT is used for each switching unit in the converter. When the DC link 

voltage is higher than 12 kV, two IGBTs in series are used for each switching unit since the required rated 

voltage is higher than 10 kV, which is the assumed maximum voltage rating for a single device.

As shown in Figure 33, the converter efficiency increases with the DC link voltage when the DC link 

voltage is lower than 12 kV. It is mainly due to two factors. Firstly, at cryogenic condition the device 

forward voltage drop is not sensitive to the amplitude of the current and the device voltage rating. As far 

as single IGBT device is used for each switching unit, higher DC link voltage will lead to lower current, 

and, therefore, lower conduction loss. Secondly, as mentioned in the previous section, the switching 

frequency will reduce if the voltage rating of the device increases. Therefore, the effect of higher 

switching loss of the high voltage rating device is compensated. When the DC link voltage is higher than 

12 kV, two IGBTs in series will be needed. The forward voltage drop for each switching unit is doubled, 

which causes the step change in the efficiency.

Figure 34 shows the weight breakdown of the converter and Figure 35 shows the overall weight 

density. As the DC link voltage increase, the size of the power stack reduces due to the improved power 

density of the higher voltage rating device. However, the size of the passive filter increases because of the 

reduced frequency. It creates a sweet spot in the voltage range of 4 to 8 kV. When the voltage crosses 

12 kV, two devices with lower voltage rating in series will be used. Here a higher switching frequency is 

feasible and leads to a step reduction in the system weight.

These results are included in the sizing tool as curve fits over the 2 to 10 kV voltage range.
NASA/CR—2015-218713 56

Figure 33.—Efficiency versus DC Link Voltage

Figure 34.—Power Converter Weight versus DC Link Voltage (Note:

“Cooling System” refers only to cryogenic pressure vessel, and not 

cryocoolers.)

2 4 6 8 10 12 14 16 18 20

0.996

0.9965

0.997

0.9975

0.998

0.9985

0.999

0.9995

Converter Efficiency vs. DC Link Voltage

DC Link Voltage (kV)

Converter Efficiency

2 4 6 8 10 12 14 16 18 20

0

100

200

300

400

500

600

DC Link Voltage (kV)

Component Weight (kg)

System Weight Breakdown

Total Weight

Power Stack

Passive Filter

Cooling System
NASA/CR—2015-218713 57

6.5 Sensitivity and Scalar Summary

6.5.1 Generators and Motors

As described above, generator and motors are scaled largely on power, not voltages. Machine design 

was not the focus of this tasked order, so recent NASA RTAPS reports were used to select mass and 

efficiency models or scalars, see Table 11.

6.5.2 Breakers and Contactors

Breaker and contactor models or scalars were developed by completing a conceptual sizing for a 

given rating and then scaling with power. The components that make up the hybrid breaker or contactor 

would all scale with power. An important assumption in the scalars below is that these components would 

be part of a pressurized system up to 650 psi. This results in a minimum mass for a pressure vessel, which 

for the AC breaker and DC contactor may be reasonable as a housing even at lower pressures. For the 

larger hybrid breaker, the pressure vessel mass may decrease if the system pressure is reduced.

Efficiency is modeled as a resistance. As reported earlier, a combination superconducting material 

and silver contact patch were proposed based on literature. It was assumed that contact resistance of 

1×10–6 Ω could be achieved for all contactors regardless of current.

6.5.3 Power Conversion

Mass and efficiency models detailed above took into consideration estimates of projected device 

ratings, switching and conduction losses, and EMI filter components, as well as a simple estimate for a 

pressure vessel containment (Table 12). From these more detailed estimates, curve fits were used to 

provide scalars or models for use at the system level (Figure 36 and Figure 37).

Figure 35.—Converter Power Density versus DC Link Voltage

TABLE 11.—GENERATOR AND MOTOR WEIGHT AND EFFICIENCY

Component Weight scalar Efficiency

(%)

Generator 78.5 kW/kg 99.955

Motor 47.1 kW/kg 99.97

2 4 6 8 10 12 14 16 18 20

22

23

24

25

26

27

28

29

Converter Power Density vs. DC Link Voltage

DC Link Voltage (kV)

Converter Power Density (kW/kg)
NASA/CR—2015-218713 58

TABLE 12.—HYBRID BREAKER AND CONTACTOR POWER DENSITY AND EFFICIENCY

Component Weight scalar Weight scalar with no 

pressure vessel

Efficiency

Hybrid AC Breaker 329 kW/kg 362 kW/kg 1×10–6 Ω per phase

Hybrid DC Breaker 139 kW/kg 188 kW/kg 1×10–6 Ω

DC Contactor 1541 kW/kg 2083 kW/kg 1×10–6 Ω

Figure 36.—Efficiency Relation Figure 37.—Power Density Relation

TABLE 13.—HTSC CORE MASS PER UNIT LENGTH AS

A FUNCTION OF DC CURRENT RATING

[+/- implies a “go and return”]

Total triax cable DC 

ampacity

(+/- kA DC)

HTSC core mass 

(kg/m)

0.9 1.28

3.8 3.51

6.0 4.74

7.5 5.59

9.0 6.45

10.2 7.04

11.8 7.39

11.8 7.39

6.6 Superconducting Cables

The mass per unit length estimates for superconducting cables are based on product data from 

transmission lines shown in Table 13. Across the range of voltages considered and the power required by 

various feeder components, an adder of approximately 4 kg/m should be used to account for the transfer 

line which carries the cryogen and provides the vacuum jacket around the HTSC core. Efficiency is 

assumed to be 100 percent besides the impact of heat leak on the cryogenic system.

y = 0.9954x0.0013

0.996

0.9965

0.997

0.9975

0.998

0.9985

0.999

0 2 4 6 8 10 12

Efficiency

Distribution Voltage (kV)

y = -0.1x2 + 1x + 25.54

25

25.5

26

26.5

27

27.5

28

28.5

0 2 4 6 8 10 12

Power Density (kW/kg)

Distribution Voltage (kV)
NASA/CR—2015-218713 59

7.0 Narrowed Range of DC Voltage

7.1 Approach to Narrowed Range of DC Voltages

With nearly all estimates based on forward-looking projections and uncertainty in the development, it 

is desirable to understand the potential impact of incorrect projects and determine if a broadly applicable 

conclusion can be reached.

Based on the work of earlier milestones, this study need only consider the power conversion 

equipment (with its impact on cryocooling capacity) and transmission lines to evaluate the impact of DC 

link voltage. Three parameters were varied over a wide range to determine if incorrect assumptions or 

estimates would greatly impact the preferred DC voltage. All cases show the greatest decrease in the 

component mass, with increasing voltage, occurs by approximately 6 kV. For the architecture selected 

and the various component assumptions, a minimum occurs at about 8 kV. For cases where cable mass 

and cryocooler mass dominate, the minimum mass is achieved at the highest voltage achievable without 

the need of paralleling two devices in a single switching unit (12 kV in this study). Though the cases with 

higher cryocooler power density may indicate a lower system mass with adoption of a 3-level power 

converter, it should be noted that this system mass is similar to the minima at a lower voltage with a 2-

level power converter. Given the wide range of estimates in length and power densities, a voltage of over 

6 kV would capture much of the weight reduction; a minimum may occur between 6 kV and 10 to 12 kV.

7.2 DC Voltage Range Selection

After more detailed consideration of all the major system components above, the sensitivities 

summarized in the last section indicate that the voltage range selection can focus on the power conversion 

equipment and transmission lines. The impact of transmission lines is simply that higher voltage allows 

smaller, lighter lines for a given system power (lower current). For power conversion, both efficiency and 

weight are sensitive to voltage. Efficiency impacts weight as the losses impact the mass of the 

cryocooling system. It should be noted that this study is assuming all components will be in a pressurized 

cryogenic environment which will also serve to provide the electrical isolation.

The relevant trends for the power electronics and transmission line masses as a function of voltage, 

for the architecture selected under this contract, are shown in Figure 38 to Figure 41. From Figure 38 and 

Figure 39, it should be noted that power electronic losses, and thus cooling system weight, decrease with 

increasing voltage. Power conversion efficiency is applied to ~100 MW, which is 50 MW generator 

power converters and 50 MW motor power converters. These losses then need to be removed by the 

cryocooler system.

In Figure 40, transmission line mass is shown to decrease with increasing voltage. In addition to the 

HTSC mass, the transfer line that carries the cryogen and provides the vacuum jacket must also be 

included. Across the range of voltages considered, this was estimated as 4 kg/m. Using these numbers and 

an HTSC line length of 700 m, the mass as a function of voltage can be estimated. This will be similar to 

the spreadsheet tool but not identical, due to the tool’s additional detail in transfer line mass estimates and 

lines of different ratings going to different components (motors and generators). Electrical insulation mass 

within feeder is minimal compared to the transfer line that provides the vacuum jacket and cryogen flow 

path.

From Table 13 and Figure 40, the reader will note that the majority of the mass is the transfer line, not 

the HTSC core. Later plots in this report will vary parameters to show trends if total transmission line 

mass per unit length were less.
NASA/CR—2015-218713 60

Figure 38.—Power Converter Efficiency vs. DC Link Voltage

Figure 39.—Power Converter Losses (for 100 MW) vs. DC Link Voltage

2 4 6 8 10 12 14

0.996

0.9965

0.997

0.9975

0.998

0.9985

Converter Efficiency vs. DC Link Voltage

DC Link Voltage (kV)

Converter Efficiency

0 2000 4000 6000 8000 10000 12000 14000

0

50

100

150

200

250

300

350

400

450

500

550

Vdc (V)

Cryocooler Mass Required by Power Electronics (kg)
NASA/CR—2015-218713 61

Figure 40.—HTSC Core and Total Transmission Line Mass (700 m) vs. DC Link Voltage

Figure 41.—Power Conversion System Mass Breakdown for 12.5 MW Generator Power Converter

0 2000 4000 6000 8000 10000 12000 14000

0

1000

2000

3000

4000

5000

6000

Vdc (V)

Mass (kg)

HTSC Transmission Line Total

HTSC core only

2 4 6 8 10 12 14

0

100

200

300

400

500

600

DC Link Voltage (kV)

Component Weight (kg)

System Weight Breakdown

Total Weight

Power Stack

Passive Filter

Pressure Vessel
NASA/CR—2015-218713 62

The trends above both point to using the highest voltage allowed by the power electronic devices in 

order to reduce system mass. The only estimate that may contribute to a system weight minima is power 

conversion mass. Specifically, the mass of filter components increases as voltage increases because 

switching frequency is estimated to need to decrease to keep switching losses reasonable. Additionally, 

the mass of the packaging or containment increases as the filter volume increases with voltage. Both of 

these influences result in increasing mass as 12 kV is approached. The containment mass (shown in 

Figure 39 and Figure 41) is for a capsule pressure vessel allowing a ~650 psi cryogen system. Plots 

shown later will use this containment or packaging mass as the extreme heavy end and will also neglect 

the containment or packaging to indicate the extreme light end of the estimate to show trends if a lower 

pressure (1 atm for example) system were selected.

The reader is reminded that the 12 kV step change (shown in Figure 38, Figure 39, and Figure 41) is 

based on a projected IGBT voltage rating beyond which a 3-level converter topology is employed. It 

should also be noted that referenced transmission cable sizing (HTSC core and transfer line) is 

extrapolated at voltages above ~10 kV.

If transmission line length is long, this will dominate weight estimates. This study estimated 700 m of 

HTSC transmission cable, including s-shaped bends between major components, and further estimated the 

need for additional cryogen transfer lines (without HTSC cores) to supply other components with 

cryogen. Earlier studies had estimated total lengths of approximately 300 m of HTSC transmission line.

Results for both lengths are considered to determine if length estimates, or equivalently mass per unit 

length, may influence voltage selection.

As previously stated, a Cryocooler mass scalar [3 kg/kW (input)] was provided by NASA for the 

purpose of converting power electronic efficiency (loss) to a mass. Figure 42 to Figure 44 show the 

impact of varying this assumption to determine its impact on the choice of DC bus voltage. This change 

in cryocooler mass scalar could equivalently be considered to show the trends of more or less efficient 

power conversion, or a higher temperature power conversion system that must not lift the heat as far.

For Figure 42 to Figure 44, the left plot does not include any packaging (pressure vessel) for the 

power converters. The right plot includes an estimate for the pressure vessel packaging of the power 

converter, which is required if the high pressure supercritical coolant system is assumed. The left and 

right plots can be compared to consider the impact of a high pressure system versus a lower pressure 

system.

The relative value of the transmission and power conversion weight represent the full system (50 MW 

generation and 50 MW fan motor loads).

Figure 43 assumes a heavier cryocooler (0.5 kW/kg), which increases the impact of power converter 

loss, thus the shape of Figure 39 is clearly visible in the figures above.

In Figure 44, the impact of power converter losses is minimized with the assumption of a very power 

dense [0.3 kg / kW (input)] cryocooler. In this plot, the shape of the power electronics mass and the 

transmission line mass trend is more visible than the shape of the power electronics efficiency trend.

In Figure 45, the cryocooler mass scalar of 3 kg/kW (input) is again used; however, a 130 K cryogen 

is assumed, which results in less cryocooler input power to lift the heat.
NASA/CR—2015-218713 63

(a) Without Power Converter Containment

(b) With Power Converter Containment

Figure 42.—Power Conversion and Cryocooling and Transmission Line Mass vs.

DC Link Voltage 3 kg/kW (input) Cryocooler scalar, 50 K cryogen, 300 K ambient

0 2000 4000 6000 8000 10000 12000 14000

0

0.5

1

1.5

2

x 104

Vdc (V)

Mass (Power Electronic + Cryo + Transmission), (kg)

300 m HTSC Transmission

700 m HTSC Transmission

0 2000 4000 6000 8000 10000 12000 14000

0

0.5

1

1.5

2

x 104

Vdc (V)

Mass (Power Electronic + Cryo + Pressure Vessel + Transmission), (kg

300 m HTSC Transmission

700 m HTSC Transmission
NASA/CR—2015-218713 64

(a) Without Power Converter Containment

(b) With Power Converter Containment

Figure 43.—Power Conversion and Cryocooling and Transmission Line Mass vs.

DC Link Voltage 15 kg/kW (input) Cryocooler scalar, 50 K cryogen, 300 K ambient

0 2000 4000 6000 8000 10000 12000 14000

0

1

2

3

4

5

6

7

8

x 104

Vdc (V)

Mass (Power Electronic + Cryo + Transmission), (kg)

300 m HTSC Transmission

700 m HTSC Transmission

0 2000 4000 6000 8000 10000 12000 14000

0

1

2

3

4

5

6

7

8

x 104

Vdc (V)

Mass (Power Electronic + Cryo + Pressure Vessel + Transmission), (kg

300 m HTSC Transmission

700 m HTSC Transmission
NASA/CR—2015-218713 65

(a) Without Power Converter Containment

(b) With Power Converter Containment

Figure 44.—Power Conversion and Cryocooling and Transmission Line Mass vs. 

DC Link Voltage 0.3 kg/kW (input) Cryocooler scalar, 50 K cryogen, 300 K ambient

0 2000 4000 6000 8000 10000 12000 14000

0

1000

2000

3000

4000

5000

6000

7000

8000

9000

10000

Vdc (V)

Mass (Power Electronic + Cryo + Transmission), (kg)

300 m HTSC Transmission

700 m HTSC Transmission

0 2000 4000 6000 8000 10000 12000 14000

0

2000

4000

6000

8000

10000

12000

Vdc (V)

Mass (Power Electronic + Cryo + Pressure Vessel + Transmission), 

300 m HTSC Transmission

700 m HTSC Transmission
NASA/CR—2015-218713 66

(a) Without Power Converter Containment

(b) With Power Converter Containment

Figure 45.—Power Conversion and Cryocooling and Transmission Line Mass vs. DC

Link Voltage 3 kg/kW (input) Cryocooler scalar, 130 K cryogen, 300 K ambient

0 2000 4000 6000 8000 10000 12000 14000

0

2000

4000

6000

8000

10000

12000

14000

Vdc (V)

Mass (Power Electronic + Cryo + Transmission), (kg)

300 m HTSC Transmission

700 m HTSC Transmission

0 2000 4000 6000 8000 10000 12000 14000

0

5000

10000

15000

Vdc (V)

Mass (Power Electronic + Cryo + Pressure Vessel + Transmission) 

300 m HTSC Transmission

700 m HTSC Transmission
NASA/CR—2015-218713 67

The summary plots above represent a broad range of density scalars. Though the absolute value and 

the existence of a minimum vary, all cases show the greatest decrease in the component mass with 

increasing voltage occurs by approximately 6 kV. For the architecture selected, and the assumptions made 

in the last milestone, a minimum occurs at about 8 to 10 kV. For cases where cable mass and cryocooler 

mass dominate, the minimum mass is achieved at the highest voltage achievable without the need to 

parallel two devices in a switching unit (12 kV in this study). Though the cases with higher cryocooler 

power density may indicate a lower system mass with adoption of a 3-level power converter, it should be 

noted that this system mass is similar to the minima at a lower voltage with a 2-level power converter.

Given the wide range of estimates in length and power densities, a voltage of over 6 kV would 

capture much of the weight reduction; a minimum may occur between 6 kV and 10 to 12 kV.
NASA/CR—2015-218713 68

8.0 Final System Weight Estimate

The final system mass roll-up was completed for an 8 kV system voltage. A high pressure (650 psi) 

cryogen system is specifically assumed in the component masses of all components, except electric 

machines, which simply reference mass scalars from earlier studies. The high pressure cryogen

assumption adds a pressure vessel mass to those components.

Figure 46 shows the evolution of the system mass estimate from the beginning of the study in the far 

left column to the final system estimate in the right two columns. Table 14 shows the estimates associated 

with the plot, and proves explanation for the weight growth. The largest weight growth is the transmission 

and distribution estimate, which now includes transfer line cryogen mass in the center column. The rightmost column includes only the HTSC core in the transmission and distribution estimate; however, the 

length of HTSC increased greatly partly due to bend radius assumptions. Power conversion mass 

increased due to a change in motor power rating and due to inclusion of a pressure vessel for the high 

pressure cryogen. In review of the figures, one can conclude that the design of the superconducting 

transfer lines and the layout of the architecture will be important to minimize system weight. Future 

systems may or may not operate at high pressure, but inclusion of some packaging mass suitable for 

structure or pressure is recommended. As was shown in the previous section, even with large variation in 

the assumptions included in this weight roll-up, a narrowed system voltage range could be projected.

A summary of the impact of a two cryogen system is provided in the following Table 15 and Table 

16. Since the majority of the electrical losses are generated by the power electronics which do not need to 

operate at temperatures as low as the superconductors, using a higher temperature cryogen may result in 

less cryocooler power and weight. Numerous system developments and details would be required for 

transitions between the two-temperature system, but the impact of the difference in required power to lift 

power conversion losses from lower temperatures could be a significant impact to system efficiency.

Figure 46.—Development of System Mass Estimates

0

2000

4000

6000

8000

10000

12000

14000

Initial Study Est. (50

MW Generation, 25

MW Fan Power)

Final Study Est. (50

MW Generation, 50

MW Fan Power)

Final Study Estimate

Est. (HTSC Core Only,

No Transfer Line

Included)

Mass (kg)

Protection

Transmission & Distribution

Power Conversion

Motors

Generators
NASA/CR—2015-218713 69

TABLE 14.—COMPONENT MASS SUMMARY

Components T&D 

subtotals 

(kg)

Final estimate Initial estimate Mass delta

(kg)

Reason for delta

Mass 

(kg)

System 

%

Mass 

(kg)

System 

%

Generators 636.9 5% 636.9 15% 0.0

Motors 1213.2 10% 606.6 14% 606.6 Due to increase in rating: Fan motor 

and generating power should be equal

Power conversion

3950.5 32% 1960 46% 718.6

Due to inclusion of pressure packaging 

and increase in rating: Fan motor and 

generating power should be equal

Transmission and 

distribution 5269.8 5269.8 43%

Ne System

HTSC 1069.8 226.8 5% 5043.0 Inclusion of transfer line estimate and 

increase in length ~330 m to 1100 m

Transfer line 3477.0

Ne cryogen 209.7

N2 System

Transfer line 432.4

N2 cryogen 80.8

Protection 1236.8 10% 867 20% 369.8 Due to fan motor rating increase and 

refinement of estimate

Total 12307 4297 8010

TABLE 15.—CRYOCOOLER POWER AND MASS ESTIMATE—TWO SEPARATE

CRYO SYSTEMS REJECTING HEAT TO AMBIENT

Ambient temperature 300 K

Supercritical Ne System Supercritical N2 System

System temperature 50 K System temperature 130 K

Carnot COP 5.0 Carnot COP 1.31

% of Carnot 30% % of Carnot 30%

Ne cooling scalar 16.7 N2 cooling scalar 4.4

Cryocooler density 3 kg/kW (input) Cryocooler density 3 kg/kW (input)

Supercritical Ne System Supercritical N2 System

Loss estimate Electrical losses Heat leak and 

pumping SC Ne

Cooling system 

input power

Heat leak and pumping 

SC LN2

Cooling system input 

power

kW % of 

System

kW % of 

System

kW % of 

System

kW % of 

System

kW % of 

System

Generators 12.55 5.7% ‐ 209.19 ‐ ‐

Motors 25.00 11.4% ‐ 416.67 ‐ ‐

Power conversion 180.78 82.8% ‐ ‐ ‐ 788.0

Transmission 0.00 0.0% 2.76 46.07 6.54 28.5

Protection 0.07 0.0% ‐ 1.23 ‐ ‐

Totals 218.40 100% 673.16 816.53

Total system losses 1708 kW 2019.5 kg 2449.6 kg

Total system efficiency 96.584%

Cryocooler mass 4469.1 kg
NASA/CR—2015-218713 70

TABLE 16.—CRYOCOOLER POWER AND MASS ESTIMATE—SINGLE

CRYOGEN SYSTEM—TEMPERATURE SET BY SUPERCONDUCTOR

Ambient temperature 300 K

Cryogen

System temperature 50 K

Carnot COP 5.0

% of Carnot 30%

Ne cooling scalar 16.7

Cryocooler density 3 kg/kW (input)

Cryogen

Loss estimate Electrical losses Heat leak and pumping Cooling system input power

kW % of system kW % of system kW % of system

Generators 12.55 5.7% ‐ 209.2

Motors 25.00 11.4% ‐ 416.7

Power conversion 180.78 82.8% ‐ 3013.0

Transmission 0.00 0.0% 5.00 83.3

Protection 0.07 0.0% ‐ 1.2

Ne System losses to N2 System ‐

Totals 218.40 100% 3723.40

Total system losses 3942 kW 11170.2 kg

Total system efficiency 92.116%

Cryocooler mass 11170.2 kg
NASA/CR—2015-218713 71

9.0 Dynamic Model and Example Case Runs

The modeling approach selected is based on Reference 45 and has been used for modeling DC power 

systems on aircraft. Large system models with detailed physics-based component models present the 

system modeler with the challenge of identifying many component parameters and typically result in 

small time constant dynamics requiring long model run times. This approach uses simplified, non-physics 

based, dynamics models. This is useful for the initial phases of system development, allowing the 

modeler to include component models that represent the dynamic performance of components while 

minimizing small time constant dynamics. Additionally, such models allow the system-level models to be 

created utilizing component models that only require top-level performance and can typically be included 

in some non-proprietary form.

9.1 Model Overview

The model assembled for this program is shown in three figures below as a roadmap to the 

Matlab/Simulink model file deliverable. Figure 47 shows the top level of the model; Figure 48 labels the 

major components; and Figure 49 labels the right, center, and left buses.

Figure 47.—Baseline System Model in Matlab/Simulink
NASA/CR—2015-218713 72

Figure 48.—Baseline System Model with Major Component Labeled

Figure 49.—Baseline System Model with Generator Bus Systems Labeled
NASA/CR—2015-218713 73

The component naming convention follows that shown in Figure 3 of the architecture definition.

The three generator buses are modeled. Connected to each generator bus are two motor feeders. The 

current from one of these two motor feeders is multiplied by one minus the total number of model feeders 

connected to the bus. This allows the model to be used for single motor feeder disconnects or faults 

without duplicating every motor feeder. Modeling every motor feeder with its associated components 

would result in longer model fun times and would not change the results, as each motor feeder is identical 

with identical parameters.

The baseline deliverable model allows investigation of nominal controls and bus stability. All 

breakers are modeled only as ideal switches; none of the energy dissipating circuitry of the hybrid 

breakers is included. This does not impact the longer time constant dynamics associated with system 

controls through fault cases; however, for fault cases it results in large voltage spikes due to forcing a 

near-instant change in current. For cases where fault voltage is the focus of the study, additional models 

were provided where only the relevant breakers were replaced with breaker models including the function 

of the energy dissipating circuitry. This can result in significantly longer model run times due to the 

addition of model states and smaller time constants. As indication of the model’s usefulness, its exercise 

for fault cases indicated an additional component, a resistor for transient damping, be added to the hybrid 

circuit breaker concept. The need for the component was made obvious when one considers system 

operation with multiple series hybrid breakers.

9.2 Component Models

Component model parameters were selected based on estimates used throughout the study effort.

These are summarized below, and are provided with full detail and comments in the model initialization 

file provided as a deliverable.

9.2.1 Generator

The main elements of the DC generator unit model are a controlled current source feeding into a 

capacitor with proportional integral feedback to control the capacitor voltage as shown in Figure 50. The 

proportional integral control regulates the generator output voltage to the set point value. The values 

Gpgen, Gigen, and Cgen can be determined from the generator and power converters’ step response, which is 

characterized by the size of the step load, the peak of the voltage transient, and the time constants of the 

step response as described in (Ref. 45).

9.2.2 Power Converter

Reference 45 represents elements of the power converter EMI filter inductance and capacitance 

parameters shown below. For this power system, the estimated EMI filter capacitance sets the capacitance 

value, but the inductance will be dominated by the series inductance in the generator hybrid DC breaker, 

which sets the inductance value shown in Figure 51.

s

Gigen −Gpgen −

Cgen

Generator

Spgen

Vgen

+

-

+

-

Figure 50.—Generator Model
NASA/CR—2015-218713 74

s

Gigen −Gpgen −

+

-

Voutgen

C

gen

Rsgen

R

pgen

Coutgen

Loutgen

Generator

I

Load

Figure 51.—Generator and Power Converter

Figure 52.—Ideal Switch Breaker Representation

The modeling approach lumps generator and power converter performance with a DC output. As a 

result of this approach, on the hybrid DC breakers are modeled; no AC transmission between generator 

and power converter is modeled.

9.2.3 Generator Breaker

When modeling to develop system controls for power sharing, system reconfiguration, and other 

slower dynamics, the breakers can be modeled as ideal switches as shown in Figure 52. Opening the ideal 

switch under load will cause voltage spikes and ringing in the system, but generally for a short time.

When investigation voltage and current during fault operation is the objective, the specific breaker 

that will open can be replaced with the model shown in Figure 53. This models the ZnO or MOV function 

of the snubber circuit as a voltage (MOV Voltage), which drives current to zero, and then the switch is 

off.

9.2.4 Contactors

For the purposes of this system model, contactors were not modeled. As the system controls are 

further developed, contactors will need added.

9.2.5 Motor

The motor incorporates an inertia parameter. This is a per unit inertia in units of seconds, meaning 

that if rated power is drawn for the defined time in seconds from an initial rated speed, the speed at the 

end of that time will be zero.

V Bus Right

+

-

VGen R Source Contactor

Cbus R

Ron

Roff

Lsource
NASA/CR—2015-218713 75

Figure 53.—Hybrid Breaker Snubber Model

9.2.6 Fan

The propulsive fans are modeled as simple cubic load as shown below.

Pload = a * speed3

The constant “a” is selected based on a fan speed and fan load. For the default model, “a” is calculated so 

that rated power of 3.57 MW is reached at 3500 rpm.

9.2.7 Bus Voltage Controls

The power converter controls and modeled generator response establish the bus voltage. The voltage 

reference is set as the nominal voltage of 8 kV, with a voltage droop offset which is used for power

sharing between generators. The voltage droop was selected as 50 V over the full power range. For a 

system with an 8 kV nominal voltage this is only 0.5 percent droop over the full power range. If it is 

determined that system disturbances cause significant variation in this range, the percentage droop could 

be increased so system disturbances have less control impact.

9.2.8 Gas Turbine Engine

The turboshaft gas turbine engine is modeled as a second order underdamped response, using the 

same approach as for the generator model. For the purpose of this simulation, a full rated power step 

results in a speed droop of 20 percent and recovery to rated speed in 5 sec.
NASA/CR—2015-218713 76

9.2.9 System Power Controls

System power is controlled by commanding fan speed. All other system components and controls 

respond to this control command, which requires increased motor current, which requires increasing 

current from the generators, and ultimately requires more torque from the engines.

9.3 Example Cases

9.3.1 Nominal Operation with Thrust Transient

The figures below show an example case of an increase in system power from a low fan speed to near 

rated conditions. The plots below are from the plotting scripts delivered with the model.

Figure 54 shows the fan load increasing with fan speed. The fan speed command was a ramp to a 

constant. All modeled fans are plotted in the windows below. The single line indicates that all modeled 

fans have exactly the same speed and load versus time. All of the following plots also plot the noted 

parameter for all components, and a single line indicates all perform the same.

Figure 55 shows engine and fan powers. The total of all 14 motor powers are applied to four 

generators and then to two engines.

Figure 56 shows the increase in engine power output and engine speed droop as the load is applied. 

Engine speed is shown to droop during the rapid increase in system load and then begin to recover as the 

increase in fan load slows. Engine response will be a combination of engine capability and controls.

Figure 57 shows the bus voltage decreasing with increasing power as defined by the droop control, 

which allows power sharing between generators. The excursion at time zero is a model initialization 

artifact.

Figure 58 shows component efficiencies, and is primarily used as a check that losses are occurring in 

the system model and they are of the approximate magnitude expected. Efficiencies will change with load 

and during transients, and could later be used to estimate the cooling capacity needed during faults where 

losses will be slightly higher. For cases where even small oscillations occur in the system, efficiency may 

be calculated to be higher than 100 percent for part of an oscillation.

Figure 54.—Fan Load and Fan Speed

0 5 10 15 20 25

0

1

2

3

x 106 Fan Load

Power (w)

Time (s)

0 5 10 15 20 25

0

1000

2000

3000

4000

Fan Speed

Speed (rpm)

Time (s)
NASA/CR—2015-218713 77

Figure 55.—Component Powers

Figure 56.—Engine Power and Turbine Speed

0 5 10 15 20 25

-4

-2

0

x 107 Engine Power

Power (w)

Time (s)

0 5 10 15 20 25

0

1

2

x 107 Generator Power (power to Generator bus)

Power (w)

Time (s)

0 5 10 15 20 25

-5

0

5

x 106 Motor Power (power from Motor bus)

Power (w)

Time (s)

0 5 10 15 20 25

0

2

4

x 106 Fan Power

Power (w)

Time (s)

0 5 10 15 20 25

-3

-2

-1

0

x 107 Engine Power

Power (w)

Time (s)

0 5 10 15 20 25

8600

8800

9000

9200

Engine Speed

Speed (rpm)

Time (s)
NASA/CR—2015-218713 78

Figure 57.—Generator and Motor Bus Voltages

Figure 58.—System Efficiencies

In addition to the system model deliverable, a model of the hybrid breaker was created using 

Matlab/Simulink SimPower components. This model allowed the confirmation of estimated hybrid 

breaker performance, but runs more slowly than elements in the larger system model. In keeping with the 

philosophy of the system model, the performance of the hybrid breaker was modeled.

0 5 10 15 20 25

7950

8000

8050

8100

8150

Generator Bus Voltages

Voltage (v)

Time (s)

0 5 10 15 20 25

7950

8000

8050

8100

8150

Motor Bus Voltages

Voltage (v)

Time (s)

0 5 10 15 20 25

99.5

100

System Efficiency

Efficiency (%)

Time (s)

0 5 10 15 20 25

99.5

100

Generator & Power Converter Efficiency

Efficiency (%)

Time (s)

0 5 10 15 20 25

99.5

100

Motor & Power Converter Efficiency

Efficiency (%)

Time (s)
NASA/CR—2015-218713 79

Figure 59.—Fan Load and Speeds for a Breaker Opening on a Single Motor

Figure 60.—Motor and Generator Bus Voltages for a Single Motor Open

9.3.2 Motor Breaker Open

This case models opening of a motor breaker under load, but not as a result of an electrical fault. Such 

operation could occur if an impending failure is detected and requires the motor bus be immediately 

disconnected from the rest of the system. The breaker must interrupt current, but not fault level currents.

Figure 59 shows a commanded ramp increase in fan speed and fan load with all fans moving together 

until time = 10 s when the motor breaker is opened. At this point, the driving torque to one fan is reduced 

0 2 4 6 8 10 12 14

0

0.5

1

1.5

2

x 106 Fan Load

Power (w)

Time (s)

0 2 4 6 8 10 12 14

0

1000

2000

3000

Fan Speed

Speed (rpm)

Time (s)

0 2 4 6 8 10 12 14

8000

8050

8100

8150

Generator Bus Voltages

Voltage (v)

Time (s)

0 2 4 6 8 10 12 14

8000

8050

8100

8150

Motor Bus Voltages

Voltage (v)

Time (s)
NASA/CR—2015-218713 80

to zero and this fan (the thin black line) slows relative to the others (thick blue line). Figure 60 shows the 

motor and generator bus voltages when the breaker opens at time = 10 s. It can be seen that no large 

voltage excursions or ringing occurs. As the load drops for one of the generator buses, the droop control 

adjusts the voltage a higher set point.

Figure 61.—Motor Bus Current During Fault (t = 10 s) and Breaker Open

Figure 62.—Hybrid Breaker Operation After Fault at t = 10 s

0 2 4 6 8 10 12

-600

-500

-400

-300

-200

-100

0

100

Current (A)

Time (s)

9.999 9.9995 10 10.0005 10.001

-600

-500

-400

-300

-200

-100

0

Current (A)

Time (s)
NASA/CR—2015-218713 81

9.3.3 Motor Fault and Breaker Open

Figure 61 shows current at the motor bus, and a current spike can be seen at time = 10 s when the 

fault occurs. On a smaller time scale, Figure 62 shows the fault current rise and the hybrid breaker driving 

current to zero.

Finally, Figure 63 shows that all of this occurs with no large voltage disturbances at the generator bus.

Figure 63.—Generator Bus Voltage During Motor Fault and Breaker Open

0 2 4 6 8 10 12

7980

8000

8020

8040

8060

8080

8100

8120

Voltage (V)

Time (s)
NASA/CR—2015-218713 82

10.0 Standards Recommendation

A good standard should be simple and of limited size, focus on the requirements formulated based on 

collective experience and consensus, utilize all relevant existing standards, and leave room for future 

improvements.

However, medium-voltage DC power systems are still not a mature technology and will require 

continued research activities. Although there are some IEEE Power Electronics Standards for Ships in 

which a MVDC power system is discussed, those standards themselves are not finalized and cannot be 

applied directly to the MVDC system under study, which is targeted for aerospace propulsion application 

and involves the use of cryogenic electrical high power equipment. Existing standards from the marine 

space are discussed in References 46 and 47. The marine space provides perhaps the closest analog, 

though it is clear that the weight sensitivity of flight will drive a different system than the marine space.

Based on the analysis of the aerospace propulsion system performed so far in this program and other 

previous work on MVDC systems, the following contents are recommended to be included in the future 

standard for MVDC power systems on aerospace propulsion systems with cryogenic electrical highpower equipment. Note that the sequence does not correspond with importance.

1. MVDC Power Ratings: The common ratings of MVDC power systems, including their operating 

devices and auxiliary equipment, should be selected from the following:

a. Rated voltage or different voltage class, which is the most important parameter of the system 

and can determine the selection and the design of the system key components, such as power 

converters, motors, cables, and generators

b. Voltage tolerance or the voltage variation range during normal operation

c. Rated withstand voltage, which includes the maximum voltage during short pulse or lightning

d. Rated continuous current, which determines the current rating of superconductor cables

e. Rated withstand current, which includes the maximum system current for short pulse, such as 

during system grounding fault

2. Grounding Method: The system components that are connected to the MVDC network, such as 

motor, generators, connectors, and power electronics converter, all need to be grounded. If not, 

the system will not have a voltage reference point and any leakage current could cause DC system 

voltage unbalance.

The grounding scheme can significantly impact the system performance, both in normal 

operating conditions and during faults. In general, solid grounding or low impedance grounding 

can introduce high leakage current during normal operation and high in-rush current, but can also 

limit system voltage; while high impedance grounding or a floating system can limit the system 

current, but can increase the system isolation requirement, especially during grounding faults.

In the standard, certain basic requirements of the system grounding method should be 

included. For example, the leakage current through the grounding network should be lower than a 

certain level.

3. Galvanic Isolation: To limit the impact of a single system fault, the whole system should be 

separated into several sections. Galvanic isolation should be provided between such sections, 

either in normal operations or at least during faults. However, galvanic isolation will also increase 

system loss, weight, volume, and cost.

In the standard, certain performance related to galvanic isolation should be addressed. For 

example, the minimum power rating of a system component or a system section should be 

identified, above which a galvanic isolation between the component and the whole system should 

be provided.
NASA/CR—2015-218713 83

4. System Stability: The MVDC system will involve many subsystems, and such subsystems need to 

be developed and tested separately before being assembled together. To guarantee a reliable 

system operation, the system stability requirements need to be included in the standard.

In the standard, a well-accepted method to describe the stability performance should be 

defined first, followed by the test methods. The minimum requirement of such performance, 

according to different operating conditions, should also be described clearly.

5. Efficiency: The electrical efficiency of the system depends on the mission of the aircraft and 

service conditions. MVDC power systems are expected to be efficient to meet economic 

constraints, especially with the use of cryogenic electrical high power equipment. Therefore, it is 

critical to achieve high efficiency under a range of operating conditions representative of 

anticipated uses. However, the use of cryogenic component also brings significant challenge into 

the cooling system, since the cooling efficiency is much lower for a cryogenic system compared 

with a system running in normal temperature. Thus, the system loss may bring more penalties for 

a cryogenic system in terms of the increased weight and cost of the cooling system.

In the standard, the lowest efficiency for each system component, such as the generator, 

motor, power converter, and cable, should be specified under different conditions, such as 

different load levels. The impact of the load variation slope should also be considered, since AC 

components can generate much more heat in a superconductor than DC. The related test methods 

for the efficiency number should also be included.

6. Quality of Service: QoS is a metric of how reliably the power system provides power to the loads.

An example of a QoS metric is the Mean-Time-Between-Service-Interruption (MTBSI). QoS 

does take into account equipment failures and normal system operation transients. Higher QoS is 

desirable. However, it usually means higher system cost.

In the standard, the minimum QoS, such as the minimum MTBSI, should be identified for 

different types of loads, e.g., the critical loads and non-critical loads.

7. Survivability: For aircraft application, system survivability is critical. Thus, in the standard the 

following topics should be addressed:

a. Fault ride-through capability. The system should be able to continue operating without being 

impacted by system faults. The system faults should be defined, as well as the severities.

Good examples include the single component failure, the system voltage drop or rise due to 

other system faults and system single-point grounding fault.

b. System redundancy. To handle the failure of certain critical components, such as the 

generator, power converter, and motor, the system design may consider involving system 

redundancy. The system redundancy may mean additional system components, such as one 

more generator, or over rating the existing system components, or both.

c. System derated operation. Under certain extreme cases, or to keep the economic benefits, the 

system should continue operating under a derated condition. The derated power should meet 

the predefined system minimum requirements.

8. Power Quality: In a DC distribution system, the fundamental frequency is zero and the concept of 

harmonic distortion in an AC system does not apply. However, the impact of the ripple voltage 

and current still exist. For the DC system, the power quality can be described by the amplitude of 

the ripple component, e.g., the maximum non-repetitive peak, the maximum repetitive peak, and 

the maximum repetitive peak-to-peak. Since in the cryogenic system the power loss in the 
NASA/CR—2015-218713 84

superconductors could be dominant by the AC or ripple currents, it is very important to cover 

power quality in the standard.

9. Operation of a Cryogenic System: Compared with a normal MVDC system, this system will 

involve the operation of a cryogenic system to enable the operation of a superconductive motor 

and generator. This special subsystem needs additional attention. In addition to providing cooling, 

the cryogenic system, at pressures above ambient, may be responsible for providing electrical 

insulation of the high voltage system.

In the standard, the performance and operation of the cryogenic system should be specified, 

e.g., the controlled temperature variation range, reliability, and the minimum cooling capability.

10. Power Management: Power management shall fulfill the following functions: manage power 

under normal conditions, maintain QoS, and maximize survivability.

Under normal operating conditions, power management must provide for on average balance 

of the energy used by loads and the energy produced by generation. In addition, power 

management also should include dynamic balancing associated with load sharing among 

generators.

Under the conditions where the power system cannot serve all loads due to damage or 

equipment failure, power management is required to implement a survivability response.

11. Others: In the standard, some other specific topics could also be addressed or reference other 

standards, e.g., the system EMI performance, system DC breaker performance, MVDC insulation 

techniques at altitude.

In addition to standards, there will be various certification requirements that will need to be 

developed with the appropriate governing bodies, such as the Federal Aviation Administration 

(FAA).
NASA/CR—2015-218713 85

11.0 Conclusions and Recommendations

11.1 Conclusions and Observations

Through investigation of several potential architectures, estimates of component weights as a function 

of voltage, comparison of electromechanical and solid-state circuit breakers, development of a dynamic 

system model, and investigation of relevant standards, the high-level conclusions and observations below 

were reached.

• A voltage source architecture was selected as the best option for reconfiguration, either for 

scalability or redundancy. Distinction between breakers (which can open under load conditions) 

and contactors (which are only opened or closed for unloaded cases) were made to reduce the 

system protection weight as contactors can be significantly lighter than breakers.

• Bus voltage in the 6 to 10 kV range was estimated to provide minimum system weight depending 

on assumptions for the forward-looking parameters. An optimum for the proposed architecture of 

approximately 8 to 10 kV was anticipated.

• Hybrid circuit breakers that use mechanical contacts for low loss current carrying, and solid-state 

components for absorption of interruption energies, were proposed and are estimated to provide 

an improved power density over mechanical breakers and improved efficiency over solid-state 

breakers.

• This range of bus voltages assumes all electrical components are in a pressurized environment to 

prevent breakdown of air at ambient pressures. The study further proposed the possibility of a 

highly pressurized system (up to ~700 psi) to allow supercritical cryogen. The potential benefit of 

supercritical cryogen is the absence of boiling of the cryogen and bubbles that concentrate electric 

fields. Such a system has the obvious drawback of requiring all component enclosures to 

withstand the cryogen pressure that contributes weight.

• Whether or not a high pressure cryogen is used, the construction of the cryogenic cables should 

be considered in weight and cooling system estimates. Existing cables with transfer line 

construction, including a flow path for the cryogen and a vacuum jacket for thermal isolation, add 

significantly to the mass of a superconductor core-only mass estimate.

• Though cooling system design was beyond the scope of this study, some estimates of cooling 

system power were needed to evaluate system losses. A simple mass scalar was also used to 

estimate the impact of cryocooler mass. As semiconductors may not require temperatures as low 

as components with superconductor material in order to achieve low losses, a two cryogen system 

may offer an option for a more efficient and lower mass system.

• The dynamic system modeling approach implemented allowed rapid assembly of a system model 

with only high-level component parameters, and was sufficient for investigating both control and 

basic fault scenarios.

• For the power and voltage levels considered, evolving standards in the marine space for Medium 

Voltage Direct Current are the closest analog and can be leveraged. These standards are, 

however, still being developed and will not likely address superconducting and cryogenic 

systems, and these standards will not likely address the weight sensitivity of systems for aircraft.

11.2 Recommendations

In general, research and development is required in all component areas to meet the mass and 

efficiency required for full turboelectric flight. System development is required to develop the power 

system architecture in coordination with the cryogenic system. Cryogenic power electronic testing and 

system developments will provide data and systems concepts to allow operating temperatures for low 

system losses when cooling is also included. System modeling for development of controls for stability 
NASA/CR—2015-218713 86

and robustness of flight-critical DC power systems will be required and could be applicable first at lower 

powers and non-cryogenic temperatures as electric flight progresses. These recommendations can be 

roughly grouped into electrical, cryogenic, and airframe-propulsion studies, and specific 

recommendations are provided below.

11.2.1 Electrical

• The DC grid stability and robustness should be further developed through modeling to investigate 

and propose local control and power sharing, as well as system management and fault 

reconfiguration approaches. The approaches would support system requirements development.

• Additionally, bus harmonics from power electronics should be studied to estimate their impact on 

the DC cables and cooling system versus filtering components.

• The cryogenic semiconductor studies completed under this effort should be furthered to include 

an improved test set-up, more devices, and a range of temperatures. This work would provide 

more recent data points than available in literature.

11.2.2 Cryogenic

• A Pressure Vessel and Cryostat Concept Trade Study for Motor / Generator / Electronics Weight 

Reduction is recommended. This trade study would evaluate the weight reduction potential of 

various cryostat concepts. Examples include (but are not limited to) strength-to-weight ratio 

enhancements like isogrid construction for metal interior pressure vessels, composite exterior 

walls with robust vacuum-tight surface treatments, honeycomb composite panel walls for 

electronics enclosures, compact rotor torque tube geometries (conical, re-entrant) with composite 

and / or hybrid metallic construction, and multi-lobed gas storage tanks.

• Concept optimization of electrical interconnection between cold neon and warm N2 flow loops is 

recommended. This trade study would evaluate and compare various material and geometry 

options for hybrid HTSC-resistive construction types versus the all-resistive aluminum alloy 

variant originally assumed for the GE baseline study. The use of HTSC (e.g., YBCO) at the cold 

Ne-anchored end of the interconnection is expected to reduce heat leak significantly due to the 

lower thermal conductivity and zero heat generation of the HTSC compared to traditional 

resistive materials used in current leads.

• Lightweight, high pressure cryogen transfer line exploration is recommended. This trade study 

aims at theoretical optimization and bench-top feasibility demonstration of a promising cryogen 

transfer line geometry and construction targeting smaller hydraulic diameter and bend radius than 

today’s commercially-available all-corrugated variants.

• AC superconductor cabling loss investigation is recommended to explore how the MgB2 cabling 

process could affect superconductor AC loss performance and critical current degradation. The key, 

high-impact, programmatic benefit of such an empirical investigation is that it would establish a 

necessary, core technical foundation leveraging subsequent machine design studies towards higher 

fidelity sizing and performance predictions. The investigation need not be machine design-specific, 

per se, but would be motor / generator-centric in that the cable forms targeted for coupon-level 

construction and testing would be amenable to machine stator slot and end-turn winding.

• LNG-based architecture studies are recommended as a parallel development path for weight 

reduction. This could compare a cryogenic system with LNG used as fuel and electrical 

component coolant to the studied superconducting systems.

11.2.3 Airframe-Propulsion

• Studies investigating fan and inlet design across aircraft size classes and mission ranges are 

recommended to understand the aircraft size and mission which could most benefit from TeDP, 

and to understand the impact on the rating and size of a TeDP system.
NASA/CR—2015-218713 87

.—Superconducting Contact

It is assumed that a 600 A, 10,000 volt superconducting contactor will be available by 2025, since 

superconducting contacts are already available in that current range. High temperature superconducting 

disconnects are presently available with current rating in the 1000s of ampere range, and the literature 

recently reports resistance in the 1 μΩ range (Ref. 32). The most promising technique at this date is 

Yttrium Boron Copper Oxide (YBCO) with a thin silver plating. Contact resistance on the order of 1 μΩ

has already been reported, with contact force in the range of 10 to 500 N, which is well within range of 

the electromechanical techniques used in conventional power contactors.

Note that “superconducting” is a misnomer for the expected contactor, since the resistance is not zero.

However, it is low enough so that the power losses (0.36 W per contactor at 600 A) are acceptable.

It is expected that the design of the superconducting contactor will resemble that of a conventional 

contactor, except its actuation coil will be superconducting, and its contacts will have very low resistance.

The size and weight of the superconducting contactor will be comparable to that of a conventional 

contactor.

The actuation coil is expected to have zero steady-state losses. It may or may not have small energy 

losses during opening and closing, but it is not expected that the contactors will be called upon to operate 

very often, so that does not really matter.

Another consideration for this application is contact life (number of switching operations). At present 

there is not much information on how many times superconducting contacts may be operated before 

requiring replacement. The technology is still in its infancy, so it is quite likely that by 2025 it will 

advance to achieve contact lifetime comparable to that of conventional contacts. Also, in this application, 

it is not expected that there will be any contact erosion due to arcing, since the contacts will be opened 

only when there is no current through them, and will be closed only when the voltage across them is zero.

Opening speed is also of interest in this application. In conventional contactors, the usual design 

employs a spring to open the contactor, and an electromagnet to close it. It is possible to increase the 

opening speed by using a spring to close, and an electromagnet to open. In any case, it is expected that 

opening time of around 1 msec is possible.

If it is required to open much faster than that, it is possible to use a repulsion disc principle. Because 

the repulsion disc would be superconducting, extremely high forces could be developed, so it would be 

possible to achieve opening times on the order of 100 μsec.

NASA/CR—2015-218713 89

.—Cryogenic Semiconductor Tests and Results

B.1 Objective

The objective of the team is to test silicon devices, both IGBT structure and MOSFET structure 

designed for high voltage operations (up to 4 kV for the IGBT and 4.5 kV for the MOSFET) at cryogenic 

temperature. The test will highlight, for the first time, the behavior of a device designed for high voltage 

operated at low temperature, and it will highlight the performances of the IGBT and MOSFET structures 

for use in high power circuitry. It must be noted that the structures are designed to work at room 

temperature and above; therefore, the team expects that the performances at cryogenic temperatures can 

be improved by designing doping profiles optimized to operate at cryogenic temperatures. The team 

cooled the devices to LN2 temperature (about 77 K) and performed switching measurements. The test is 

aimed to establish the conduction and switching losses of high voltage rated devices in cryogenic 

environment.

B.2 Introduction

Two devices have been considered for the cryogenic tests: the IGBT IXEL40N400 and the MOSFET 

IXTL2N450, both of them produced by IXYS. These devices have been considered as an example of high 

voltage silicon parts that could be used in a high power application.

The IGBT is rated for 4 kV and 90 A at 25 °C (it decreases to 40 A if the junction temperature is 

110 °C), and the MOSFET is rated for 4.5 kV and 2 A at 25 °C (and a differential resistance of 20 Ω 

approximately). Both parts have and identical footprint, and both parts use the same amount of silicon: 

14.26- by 14.26- by 0.6-mm approximately.

Figure 64 shows the IGBT (on the left) and the MOSFET (on the right) in their commercial package 

(bottom) and without the plastic envelop (top). Figure 65 shows the active silicon area completely isolated 

and overlapped on the devices (IGBT on the left and MOSFET on the right), and Figure 66 shows the two 

active areas next to each other (IGBT on the left and MOSFET on the right), showing that the two devices 

use the same identical area.

Figure 64.—IGBT (left) and MOSFET (right) Full 

Package and Stripped Version

Figure 65.—IGBT (left) and MOSFET (right)

Full Package Overlapped With Active Silicon Area
NASA/CR—2015-218713 90

Figure 66.—Active Silicon Area of IGBT (right) and MOSFET (left)

Figure 67.—Resistive Switching Schematic

The authors consider the comparison of the two structures a fair comparison based on the fact that 

they use the same silicon area.

B.3 Test Set-Up

The test set-up follows a similar procedure as the one listed in (A. Caiafa 2004). The testing circuit 

with resistive load is represented in the schematic shown in Figure 67. While the symbol of the device in 

Figure 67 is relative to the IGBT, the same circuit has been used for both IGBT and MOSFET switching 

testing.

The test bed was assembled to minimize the parasitic inductance and to minimize the thermal losses 

such that the temperature of the device can be considered constant throughout the test. In order to 

minimize parasitic inductance, coaxial cables were used for every connection from inside the cryostat to 

outside, and to minimize thermal losses, thermal barriers were added to the set-up. The thermal stability 

of the device was also greatly improved by the use of a large heat sink. Figure 68 shows the set-up before 
NASA/CR—2015-218713 91

being inserted into the cryostat: the large heat-sink is visible on the bottom, the device is mounted on the 

heat-sink and it is connected through four coaxial cables to the top of the insert. In Figure 68, the two 

thermal barriers are also visible. The team used four coaxial cables to implement the four probe 

measurement technique for both the conduction (two coaxial cables used for a total of four conductors) 

path and the gate controls (two coaxial cables used for a total of four conductors).

Figure 69 shows the full set-up that includes: the LN2 tank, the test cryostat, the oscilloscope, and the 

power supply. Not visible in the figure is the resistive load and the capacitor bank (behind the 

oscilloscope).

The insert was equipped with thermal probes as well: one next to the device, one below the heat-sink, 

and one a few centimeters above the device. The last probe was mainly used to make sure that the device 

is fully covered in LN2 during the test.

Figure 68.—Test Insert

Figure 69.—Test Set-Up
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B.4 Static Tests

The team has tested both IGBT and MOSFET, collecting the V-I curves at cryogenic temperature.

After slowly cooling the device, once the device is completely submerged by LN2 and has been 

submerged for more than one minute, the device has been collected to a high power curve tracer and the 

V-I characteristic was collected. The test has been repeated two times per device. Only one IGBT and one 

MOSFET were tested.

Thanks to the high power curve tracer, the team was able to test the IGBT with currents as high as 

200 A. Figure 70 shows the V-I characteristic of the IGBT. Focusing first on the characteristic at LNT 

and room temperature (25 °C), it is possible to observe that the IGBT at LNT has an improvement of 

about 38 percent.

It must be stated that the thermal resistance of the IGBT packaging decreases by about 5 to 10 times 

compared to the room temperature thermal resistance. In other words, losses that will increase the 

temperature of the junction of the IGBT of 10 °C, for example, will cause an increase of junction 

temperature of 50 to 100 °C in an IGBT kept to 25 °C; therefore, it is more meaningful to compare LNT 

performances to 125 °C junction performances. Comparing these two characteristics, shown in Figure 70

in solid blue (LNT) and in red (125 °C), it is possible to observe that the losses are about 55 percent less 

at LNT.

Figure 70.—V-I Characteristic of the IGBT. Measured at LNT (solid blue), two IGBTs in parallel at LNT (marked 

as 2p IGBT), 6 IGBTs in parallel at LNT (marked as 6p IGBT). Characteristic at room temperature (yellow 

trace), and at 125 °C (red trace).
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If lower conductive voltage drop is needed, more devices can be paralleled. In Figure 70 the curves of 

two IGBTs paralleled and six IGBTs paralleled are shown. While the overall voltage drop does not 

decrease in proportion, a much larger current can be conducted for a relatively small junction temperature 

increase. Paralleling more than six devices becomes quite difficult due to device characteristic pairing and 

parasitic inductance issues.

Figure 71 shows the V-I characteristic of the MOSFET at LNT, room temperature, and 125 °C.

Figure 71 shows that a cryocooled MOSFET has 80 percent less losses than the same MOSFET 

operated at room temperature, and about 90 percent less losses than a MOSFET with a junction 

temperature of about 125 °C (about 10X improvement). As with the IGBT, the thermal resistance 

drastically decreases at LNT, making it more meaningful to compare the LNT characteristic with the 

125 °C characteristic than comparing it to the 25 °C characteristic.

In summary, the cryocooled IGBT exhibits a 2 times improvement, while the cryocooled MOSFET 

exhibits a 10 times improvement in conduction losses performances.

Despite the fact that the MOSFET has a much larger improvement than the IGBT, the IGBT has a 

much better power density capability. Figure 72 shows a comparison of the cryocooled IGBT and 

MOSFET conduction performances. It is clear that, given the same silicon area and same voltage 

blocking capability, the IGBT can conduct a much larger current. In order to match the performances of 

the IGBT at 200 A, about 160 MOSFETs are required to be connected in parallel.

Figure 71.—V-I Characteristic of the MOSFET. At LNT (blue trace), at room temperature (yellow trace), 

and at 125 °C (red trace)
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Figure 72.—Comparison of the V-I Characteristics of the IGBT (blue trace) and 

MOSFET (red trace). Measured at LNT

The breakdown voltage of the IGBT at LNT decreased to 3.6 kV, while the breakdown of the 

MOSFET decreased to 3.2 kV. It must be noted that only one IGBT and one MOSFET have been tested.

To have a more statistically meaningful value, a larger number of devices need to be tested.

In conclusion, a MOSFET structure exhibits a larger improvement, but an IGBT structure still 

exhibits much less losses for same silicon area and voltage blocking capability.

B.5 Dynamic Testing

B.5.1 Resistive Test Results

The resistive test was performed using a low inductance high energy dissipation resistor of 

approximately 5 Ω. The voltage drop on the device was measured through a voltage divider and a zener 

diode that clamps the measured voltage when this rises above the zener voltage (in this experiment it was 

about 15 V).

An example of the data collected during the resistive switching test is shown in Figure 73. In this 

figure, the green trace corresponds to the current through the Device Under Test (DUT), the red trace 

corresponds to the voltage drop across the DUT, and the purple trace corresponds to the capacitor voltage.

The controller used to control the circuit provides a double pulse command; while this is not necessary for 

a resistive switching, it will be for the inductive switching. The team used this data to measure the voltage 

droop at various currents. The challenge in this measurement is the transition time (visible in the red 

trace) due to the use of the zener and the voltage divider, and the accuracy of the Vce (collector-emitter 

voltage during conduction) measurement. It is quite difficult to distinguish the voltage droop of the device 

from the zener transition.
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Figure 73.—Oscilloscope Traces for Resistive Test at 40 A

Figure 74.—First Set of Data Captured at LN2 Temperature

Figure 74 shows a set of data collected using the set-up and measurement system described above. 

While the results of the resistive dynamic measurements at cryogenic temperature are quite comparable 

with the data collected statically (V-I curves), the team feels that the resistive dynamic measurement setup needs improvement and is working toward it.
NASA/CR—2015-218713 96

B.5.2 Inductive Test

The team has prepared the inductive load and the test bed for the inductive switching as shown in 

Figure 75. The inductor built is 510 μH and the recirculating diode is a Schottky SiC diode rated for 

100 A. The selected diode will minimize the reverse current effect during the turn on, allowing the team 

to perform more accurate losses evaluation.

Figure 76 shows double pulse waveforms collected using a cryocooled IGBT. The purple trace 

represents the voltage and the green trace represents the current. Figure 77 and Figure 78 show the turn on 

and turn off, respectively.

Figure 75.—Inductive Switching Test Set-up
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Figure 76.—Detail of the Double Pulse Test of the Inductive Switching at LNT

Figure 77.—Inductive Turn-On at LNT
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Figure 78.—Inductive Turn-Off at LNT

The set-up was operated to produce 600 V and 30 A. The initial objective was to push the device at 

higher voltage and higher current, but, because of the excessive current ringing (both overshoot and 

undershoot), the team had to limit the test to the mentioned value. The large current peaks can damage not 

only the device under test, but especially the equipment connected around it. The current overshoots are 

caused by the excessive parasitic inductance, measured to be around 500 Nh, that is introduced by the 

relatively long coaxial cables and by the very fast turn on and turn off speed of the DUT. The long coaxial 

cables were needed to lower the DUT into the cryogenic bath in the available cryo-tank. The very fast 

speed of the device is due to the increase in the carrier mobility at cryo temperature.

By observing the curves shown in Figure 77 and Figure 78, we can argue that the device switches in 

an extremely short time (the current reaches the maximum peak within 10s of nanoseconds), but, because 

of the large ringing caused by the parasitic inductance, not much more can be said.

The MOSFET is expected to switch faster than the IGBT, so the issues caused by the parasitic 

inductance will be magnified. This is the reason why the team decided not to test the MOSFET within this 

set-up.

B.6 Conclusions

The team proved that there are benefits in using cryocooled large voltage blocking devices. A more 

detailed study is needed to completely assess the benefit of high voltage cryocooled devices (new 

dynamic test bed).

A different cryo-tank has been identified that will require much shorter coaxial cables; therefore, a 

much lower parasitic inductance, to repeat the dynamic tests and complete the assessment of the two DUT 

(IGBT and MOSFET).
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Given the results obtained, the team feels confident that a system based on the use of superconductors 

and cryo-power electronics will be beneficial to the overall system efficiency, especially if the system 

utilizes a “disposable” coolant such as a fuel. A two temperature cryogenic system (operating at two 

different cryo temperatures) may be the ideal choice.

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.—Model Description

The file below provides the parameters used for the Matlab/Simulink system model used for this 

program.
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