Michael J. Armstrong and Mark Blackwelder

Rolls-Royce North American Technologies, Inc. (LibertyWorks), Indianapolis, Indiana

Andrew Bollman and Christine Ross

Rolls-Royce Corporation, Indianapolis, Indiana

Angela Campbell

Georgia Institute of Technology, Atlanta, Georgia 

Catherine Jones and Patrick Norman

University of Strathclyde, Glasgow, Scotland, United Kingdom

Architecture, Voltage, and Components for a 

Turboelectric Distributed Propulsion Electric Grid

Final Report

NASA/CR—2015-218440

July 2015

EDNS04000038188/002
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Michael J. Armstrong and Mark Blackwelder

Rolls-Royce North American Technologies, Inc. (LibertyWorks), Indianapolis, Indiana

Andrew Bollman and Christine Ross

Rolls-Royce Corporation, Indianapolis, Indiana

Angela Campbell

Georgia Institute of Technology, Atlanta, Georgia 

Catherine Jones and Patrick Norman

University of Strathclyde, Glasgow, Scotland, United Kingdom

Architecture, Voltage, and Components for a 

Turboelectric Distributed Propulsion Electric Grid

Final Report

NASA/CR—2015-218440

July 2015

EDNS04000038188/002

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NASA/CR—2015-218440 iii

Contents

1.0 Introduction ........................................................................................................................................... 1

1.1 TeDP Electrical System ................................................................................................................ 1

1.2 A Review of Current Voltage Standards ....................................................................................... 2

1.2.1 Grid Codes ........................................................................................................................ 4

1.2.2 Maritime Power Systems .................................................................................................. 8

1.2.3 Current Aircraft Voltage Standards ................................................................................ 11

1.3 Challenging in Creating DC Propulsion Standards ..................................................................... 12

1.4 TeDP Electrical System Voltage Standards ................................................................................ 13

1.4.1 Regulation, Protection, and Recovery: A Design/Engineering Perspective ................... 13

1.4.2 Operational Voltage Limits Category Definitions .......................................................... 14

1.4.3 Selection of the Optimal Operating Voltage ................................................................... 16

1.5 Introduction Summary ................................................................................................................ 19

2.0 Architecture Selection ......................................................................................................................... 20

2.1 Architecture Candidates .............................................................................................................. 20

2.1.1 Concept 1: Baseline Architecture.................................................................................... 20

2.1.2 Concept 2: Inner Bus Tie Concept .................................................................................. 21

2.1.3 Concept 3: 3-Bus Multifeeder Concept ........................................................................... 21

2.1.4 Concept 4: Cross-Redundant Multifeeder Concept ........................................................ 21

2.1.5 Concept 5: 4-Bus Inner Bus Tie Multifeeder Concept .................................................... 22

2.2 Architecture Evaluation and Selection ........................................................................................ 24

2.3 Weight Sensitivity and Deliverable Objectives .......................................................................... 25

2.4 Overview and Naming Convention for Selected TeDP Architecture ......................................... 27

3.0 Terrestrial Systems Benchmarking ..................................................................................................... 29

3.1 Power Transmission Superconducting Cable Installations ......................................................... 29

3.2 Future DC Power Transmission Installations ............................................................................. 30

3.3 Superconducting Fault-Current Limiter Installations and Prototypes......................................... 31

3.4 High Power Normally Conducting Solid-State Switchgear ........................................................ 31

3.5 Cryogenic Semiconductors ......................................................................................................... 32

3.6 Cryogenic Power Converter Prototypes ...................................................................................... 34

3.7 Superconducting Electric Machine Prototypes ........................................................................... 35

3.8 Study Voltage Range Conclusion ............................................................................................... 35

4.0 DC Protection Devices ........................................................................................................................ 37

4.1 Introduction ................................................................................................................................. 37

4.2 DC Electromechanical Circuit Breakers ..................................................................................... 38

4.2.1 Conventional DC EMCBs ............................................................................................... 38

4.2.2 EMCBs Applied to Superconducting Systems ............................................................... 39

4.3 Hybrid Circuit Breakers .............................................................................................................. 42

4.4 Solid-State Circuit Breakers........................................................................................................ 45

4.5 Summary ..................................................................................................................................... 47

5.0 Component Sensitivities and Sensitivity Modeling ............................................................................ 48

5.1 Component Sensitivity Overview ............................................................................................... 48

5.2 Rectifier and Inverter .................................................................................................................. 51

5.2.1 Model Overview ............................................................................................................. 52

5.2.2 Source-Side Converters ................................................................................................... 54

5.2.3 Source-Side Converters Trends....................................................................................... 67

5.2.4 Load-Side Converters ..................................................................................................... 73

5.2.5 Load-Side Converter Trends ........................................................................................... 75

5.3 Cables Model Overview.............................................................................................................. 80

5.3.1 Cable Layout ................................................................................................................... 80
NASA/CR—2015-218440 iv

5.3.2 Monopolar, Bipolar, Redundancy and Three Phase AC ................................................. 81

5.3.3 Superconducting Material ............................................................................................... 82

5.3.4 Applied Field .................................................................................................................. 83

5.3.5 Cryostat ........................................................................................................................... 83

5.3.6 Cooling ............................................................................................................................ 85

5.3.7 Losses .............................................................................................................................. 85

5.3.8 AC Losses ....................................................................................................................... 86

5.3.9 Dielectric ......................................................................................................................... 86

5.3.10 Mass and Efficiency Trends ............................................................................................ 87

5.3.11 DC Cable ......................................................................................................................... 89

5.3.12 AC Cable ......................................................................................................................... 90

5.3.13 Cable Losses ................................................................................................................... 91

5.4 Superconducting Magnetic Energy Storage ................................................................................ 92

5.4.1 Parameter Diagram ......................................................................................................... 93

5.4.2 Sizing Equations ............................................................................................................. 94

5.4.3 Power Electronics ........................................................................................................... 97

5.4.4 Cryocooling Weight Calculations ................................................................................... 98

5.4.5 SMES Sensitivity Study Parameters and Assumption Overview ................................... 98

5.4.6 Mass and Efficiency Trends ............................................................................................ 99

5.5 Circuit Breakers ........................................................................................................................ 105

5.5.1 Model Overview ........................................................................................................... 106

5.5.2 Mass and Efficiency Trends .......................................................................................... 108

5.6 Superconducting Fault Current Limiters ................................................................................... 116

5.6.1 Modeling Approach ...................................................................................................... 116

5.6.2 Geometry and Winding ................................................................................................. 117

5.6.3 Parameter Diagram ....................................................................................................... 119

5.6.4 Governing Equations..................................................................................................... 120

5.6.5 Quench Modeling .......................................................................................................... 120

5.6.6 Quench Cooling ............................................................................................................ 121

5.6.7 Induction Equations ...................................................................................................... 122

5.6.8 Mass Equations ............................................................................................................. 124

5.6.9 Assumptions Overview ................................................................................................. 124

5.6.10 Geometry Sensitivity..................................................................................................... 129

5.6.11 Efficiency Trends .......................................................................................................... 131

5.7 Generators ................................................................................................................................. 134

5.8 Cryogenic System ..................................................................................................................... 134

5.9 Component Sensitivity Modeling Summary ............................................................................. 134

6.0 Narrowed DC Voltage Range............................................................................................................ 135

6.1 Introduction ............................................................................................................................... 135

6.2 Architecture Assumptions ......................................................................................................... 135

6.3 Electrical System Mass Sensitivity to Voltage ......................................................................... 137

6.4 Electrical System Cooling Requirements Sensitivity to Voltage .............................................. 139

6.5 IGBT and Diode Switching and Conduction Losses ................................................................ 142

6.6 Cryocooler Mass ....................................................................................................................... 142

6.7 Narrowed Voltage Range .......................................................................................................... 144

7.0 GT Dynamic Model .......................................................................................................................... 147

7.1 Introduction ............................................................................................................................... 147

7.2 Superconducting Fault Current Limiter .................................................................................... 147

7.2.1 SFCL Modeling Overview ............................................................................................ 147

7.2.2 SFCL SimPowerSystems Model ................................................................................... 148

7.2.3 SFCL State-Space Model .............................................................................................. 153
NASA/CR—2015-218440 v 

7.3 Solid-State Circuit Breaker ....................................................................................................... 154

7.3.1 SSCB Modeling Overview ............................................................................................ 155

7.3.2 SSCB SimPowerSystems Model .................................................................................. 157

7.4 Energy Storage Model (SMES) ................................................................................................ 159

7.5 Power Converter Models .......................................................................................................... 161

7.5.1 Rectifiers ....................................................................................................................... 162

7.5.2 Inverters ........................................................................................................................ 172

7.6 System Modeling ...................................................................................................................... 177

7.6.1 Single Motor Model ...................................................................................................... 178

7.6.2 Fault Isolation Model .................................................................................................... 182

7.6.3 Nominal Recovery Model ............................................................................................. 182

7.6.4 Failure Scenarios ........................................................................................................... 182

7.6.5 Running the Models ...................................................................................................... 195

7.7 Summary and Future Studies .................................................................................................... 196

8.0 Conclusions ....................................................................................................................................... 197

8.1 Power Electronics ..................................................................................................................... 197

8.2 Protection .................................................................................................................................. 198

8.3 Energy Storage .......................................................................................................................... 198

8.4 Distribution ............................................................................................................................... 199

8.5 Dynamic Modeling ................................................................................................................... 199

Appendix A.—Acronyms and Abbreviations ........................................................................................... 201

Appendix B.—IGBT Data ........................................................................................................................ 203

Appendix C.—Dynamic Models .............................................................................................................. 205

Appendix D.—Strathclyde Report ............................................................................................................ 213

References ................................................................................................................................................. 252

NASA/CR—2015-218440 1 

Architecture, Voltage, and Components for a Turboelectric 

Distributed Propulsion Electric Grid

Final Report

Michael J. Armstrong and Mark Blackwelder

Rolls-Royce North American Technologies, Inc. (LibertyWorks) 

Indianapolis, Indiana 46207

Andrew Bollman and Christine Ross 

Rolls-Royce Corporation 

Indianapolis, Indiana 46206

Angela Campbell

Georgia Institute of Technology

Atlanta, Georgia 30332 

Catherine Jones and Patrick Norman

University of Strathclyde 

Glasgow, Scotland, G1 1XQ, United Kingdom 

1.0 Introduction

1.1 TeDP Electrical System

The development of a wholly superconducting turboelectric distributed propulsion system presents 

unique opportunities for the aerospace industry. However, this transition from normally conducting 

systems to superconducting systems significantly increases the equipment complexity necessary to 

manage the electrical power systems. Due to the low technology readiness level (TRL) nature of all 

components and systems, current Turboelectric Distributed Propulsion (TeDP) technology developments 

are driven by an ambiguous set of system-level electrical integration standards for an airborne microgrid 

system (Figure 1). 

While multiple decades’ worth of advancements are still required for concept realization, current 

system-level studies are necessary to focus the technology development, target specific technological 

shortcomings, and enable accurate prediction of concept feasibility and viability. An understanding of the 

performance sensitivity to operating voltages and an early definition of advantageous voltage regulation 

standards for unconventional airborne microgrids will allow for more accurate targeting of technology 

development. 

Propulsive power-rated microgrid systems necessitate the introduction of new aircraft distribution 

system voltage standards. All protection, distribution, control, power conversion, generation, and 

cryocooling equipment are affected by voltage regulation standards. Information on the desired operating 

voltage and voltage regulation is required to determine nominal and maximum currents for sizing 

distribution and fault isolation equipment, developing machine topologies and machine controls, and the 

physical attributes of all component shielding and insulation. Voltage impacts many components and 

system performance. 
NASA/CR—2015-218440 2 

Figure 1.—Rolls-Royce TeDP System Architecture Concept. 

Issues that must be considered for higher voltage operation are the following: 

• Maximum and steady-state current ratings 

• Fault current interruption for fault isolation and protection 

• Partial discharge and corona protection

• Dielectric insulation life

• Safety procedures for testing, manufacturing, and maintenance

To assess the feasibility of unconventional airborne N+2/3 microgrid concepts more accurately, tools 

and processes are necessary to estimate and govern the development of appropriate voltage regulation 

standards for these unconventional concepts. An integrated system-level process for identifying the 

desired voltage standards would allow for more accurate prediction of electrical system weight and 

volumes, overall system reliability, and safety of technology concept. This influences the vehicle-level 

feasibilities assessments, highlighting technology gaps where increased development is needed 

(protection equipment). Once advantageous sets of standards are identified, these standards can be used to 

assist in informing electrical technology development relating to expected systems interactions and 

operational constraints. 

A holistic approach to defining the appropriate standards is necessary to guide future technology 

developments. Most current superconducting technology developments are subject to constraints imposed 

by interfacing with normally conducting terrestrial systems. However, because the NASA N3-X 

microgrid is wholly superconducting, many of the driving connection constraints are removed, and the 

operating standards for a wholly superconducting system can be tailored to capture maximum benefit to 

the entire system. Additionally, because this system must be airworthy and flight critical, operating 

standards must be defined that lend specifically to aircraft environmental, safety, and performance 

objectives.

1.2 A Review of Current Voltage Standards

Terrestrial systems have adopted multi-kilovolt (kV) standards for electrical power distribution. 

However, the aerospace community typically operates well under the kV level. The highest accepted 

power distribution voltage for conventional transport aircraft is ±270 Vdc. These common voltage 

practices reflect the current aircraft electrical power systems paradigm. While increased voltages would 
NASA/CR—2015-218440 3 

act to reduce conductor weight and volume, insufficiently understood risks associated with insulation and 

protection equipment prohibit the introduction of higher voltage standards. 

Voltage levels on existing electrical systems for aircraft are relatively low. As power demands 

increases on a conventional system, it is necessary to increase voltage levels so that conductor weight can 

be reduced. Increasing the voltage level allows the conductor current to decrease for the same power 

requirement. Reduced conductor current rating also reduces conductor weight. However, higher voltage 

levels require thicker insulation, which contributes to an increase in cable weight. 

The primary motivation for current limitation on voltage for airborne power system is derived from 

Paschen’s Law (Figure 2) (Ref. 1). This law considers parallel metal plated in air under a uniform electric 

field. Figure 2 charts the voltage breakdown of an airgap in terms of the product of the distance between 

conductors and pressure. The approximate minimum breakdown voltage for any product of pressure and 

distance is 327 Vdc. This means an arc will not occur between two parallel metal plates at voltage levels 

below this value at low or high altitude. For this reason, existing aircraft dc voltages remain below the 

327-Vdc threshold. 

The superconducting cables and physical layout of the electrical system must consider Paschen’s Law 

to avoid breakdown and discharge especially at high altitude by designing cable insulation and distance 

between conducting elements appropriately. In addition to the composition of the airgap, pressure, and 

conductor distance, the breakdown threshold voltage is also a function of temperature. For lower 

temperatures, which increase the air density, the strength of the gap is increased since increase in air 

density translates to a larger pressure-distance product. Since this electrical system is cryogenic, the 

breakdown voltage may be less sensitive to pressure and conductor distance than the room-temperature 

curve for which Paschen’s Law is characterized. Other factors that should be considered when 

determining system voltage levels are contamination in the air gap, impact of vibration, abnormal system 

events, and transient events (Ref. 2).

Conventional wisdom indicates that increased power levels demand higher voltages to reduce 

operating current and thus reduce the conductor size and weight. However, in the context of a wholly 

superconducting microgrid system, overall system weight becomes largely insensitive to increases in 

conductor cross-sectional area. Superconducting components have acceptable power density with lowervoltage higher-current levels. An additional challenge arises, however, for large current systems at lower 

voltages. In this case, large operating currents require larger interruption devices for protection and 

control purposes. 

Figure 2.—Illustration of Paschen’s Law (Ref. 1).
NASA/CR—2015-218440 4 

Extrapolations from terrestrial superconducting standards provide a limited basis for comparison due 

to typical fundamental assumptions applied. Current implementations of superconducting equipment are 

developed and integrated in a piecemeal fashion. Sections of superconducting cabling or machine 

equipment are coupled with normally conductive systems for specific targeted benefits. In these cases, the 

weight, size, or cost of a normally conductive component becomes prohibitive and benefits to the 

insertion of the superconducting component warrants insertion into the more conventional system. As 

such, the operating requirements for the superconducting component are fixed by the bounding systems 

elements or defined by considering the single component’s performance sensitivity in isolation. 

1.2.1 Grid Codes

Voltage standards that apply to the utilities industry vary according to country and region. Two areas 

reviewed here are the North American market and the European market. The North American grid is stiff 

but is starting to develop standards for the connection of small, distributed generation to the larger power 

system. The European market is more dynamic and the standards take into account the integration of 

disruptive power sources such as photovoltaic (PV) systems. The utilities’ standards normally assume that 

the power on the grid is shared by multiple generators. While this sharing of power on a common grid is 

dissimilar to the TeDP project, some trends are worth noting.

1.2.1.1 European Standards

The British Standards Institute (BSI) released requirements for voltage regulation and parallel operation 

of AC machines (Ref. 3). BS 4999-140 covers generators running in parallel and running singly. This 

standard is unique as it uses a percent of rated current to define load changes rather than rated power and 

faulted conditions are not covered in the standard. The standard defines nine regulation grades that a 

generator can be specified to depending on the application and agreement between manufacturer and 

purchaser. Permissible steady-state voltage regulation, transient voltage droop, and recovery times are given 

for each regulation grade. The amount of load acceptance can vary from 35 to 100 percent of rated current, 

at rated voltage, and at a power factor between 0.4 and 0 lagging. For load shed events, the amount of 

allowable for voltage droop is given for 100 percent load shed events assuming a 0.8 power factor.

The bulk European energy market is not an open market between countries, so while the International 

Electrotechnical Commission (IEC) provides some overarching regulations, such as standard voltages

(Ref. 4), individual countries impose additional regulations on their utility companies. It is also worth 

noting that because of this fragmentation, the European electric grid is not as “stiff” as the North 

American grid, so it may provide better insight on how disruptive technologies (wind, photovoltaic 

penetration) impact system regulation. In the Finnish specifications for power plant performance (Ref. 5), 

the nominal voltage in the main grid can drop to as low as 0.85 pu and up to 1.05 pu during disturbances, 

while the nominal voltage is within 0.95 and 1.05 pu. Unlike aircraft standards, the voltage and frequency 

regulations are dependent on each other as Figure 3 indicates. Generators may remain synchronized to the 

grid as long as they do not suffer damage if the frequency is below 47.5 Hz or above 53 Hz. Generators 

must be disconnected if the frequency is above 55 Hz. 

For disturbances and exceptional conditions, the generating units are designed to withstand the 

suggested grid voltage variations shown in Figure 4 without disconnection from the grid and only a small 

power reduction is accepted.

The Finnish grid code further states the preference to have generators with low reactances and also 

specifies the amount of forcing the exciter is expected to have so that generator can ride through transient 

conditions. Each generator is capable of operating at the rated active power continuously with a power factor 

down to at least 0.95 underexcited and 0.9 overexcited, throughout the voltage rant of 100 to 105 percent on 
NASA/CR—2015-218440 5 

the underexcited condition and 90 to 105 percent on the overexcited condition. The voltage control system 

includes a power system stabilizer (PSS), protective limiters and reactive current statics equipment.

A second example of voltage control is the new German grid codes for connecting photovoltaic 

systems to medium voltage power grids in accordance with DIN EN 50160 – “Voltage Characteristics of 

electricity supplied by public distribution networks” (Refs. 6 and 7). This is relevant as the grid considers 

power connected to the grid through converters. 

Figure 3.—Finnish Grid Requirements concerning Power 

Production When the Grid Frequency and Voltage Vary. 

Figure 4.—Grid Voltage Transient Caused by a Fault. 
NASA/CR—2015-218440 6 

Figure 5 shows the limiting curves during a fault for plants connected to the grid that do not use a 

synchronous generator. It can be seen that the power plant must remain tied to the grid during a voltage 

drop down to 0 V for up to 150 ms. If voltage is above boundary line 1, then the unit has to remain 

connected to the grid. If the voltage is between boundary line 1 and boundary two, the behavior of the 

system can vary based upon agreement between network operator and plant operations. Between 

boundary line 2 and the blue line, disconnect times greater than 2 s are allowed to protect generation 

equipment subject to agreement with the network operator. 

In the event of a network symmetrical fault that results in weakened voltage, the power facility must 

manage the import and export of reactive power to prevent a full voltage collapse. Figure 6 shows the 

voltage response during a symmetrical fault so that the generator injects reactive current onto the network. 

The generating units must be capable of feeding the required reactive current into the grid within 20 ms of 

the fault, up to 100 percent of the rated current. The German standard also contains a power reduction 

formula similar to the Finnish grid study above that helps to protect the equipment from unsafe system 

operation.

Figure 5.—Limiting Voltage Curves at the Grid Connection Point in the Event of a Network Fault. 

Figure 6.—Principle of Voltage Support in the Event of a Symmetrical Network Fault. 
NASA/CR—2015-218440 7 

1.2.1.2 North American Standards

There are numerous standards that cover the electric grid in the United States, from the federal 

regulations of the North American Electric Reliability Corporation (NERC) to the regional Independent 

Service Operators (ISO) to the industry standards of IEEE 1547 and ANSI C84. There are standards that 

cover the generation, transmission, distribution and consumption of electrical energy based on a 

generator’s size, its location and if it is participating in the wholesale energy market. This review only 

covers a very small section of the standards.

The NERC ensures the reliability of the bulk power system in North America. The standard NERCPRC-024 relates to the frequency and voltage protective settings of generators and ensures that generators 

remain connected during defined frequency and voltage excursions. PRC-024 defines the voltage during a 

fault at the Point of Interconnection (POI). Transmission connection systems (large utilities) have a set of 

different requirements than DER including voltage tolerance in accordance with NERC PRC-024 and 

allow the voltage at the source to be controlled on voltage, power factor or reactive power. 

IEEE 1547 is the standard for interconnecting Distributed Energy Resources (DERs) with electric 

power systems. Table 1 shows the voltage and frequency variations allowed by IEEE 1547. The standards 

cover the interconnection of DER by imposing requirements on certain voltage characteristics (Ref. 8). 

Voltage regulation is to be within ±5 percent per ANSI C84, voltage control of DER is not permitted per 

IEEE 1547, the distortion of the waveform is controlled by individual harmonics, and the IEEE 1547 

requires that the direct current injection in the grid be less than 0.5 percent of the full rated RMS output 

current. 

Figure 7 superimposes the transient voltage curves of both the transmission system (NERC PRC-024) 

and the distributed system (IEEE 1547). The curves look similar to the European fault voltage curves as 

well as the voltage envelopes of MIL-STND-704. This does not dictate the voltages of the system or the 

voltage at the load but regulates voltage at the point of connecting an electrical source to the grid. 

TABLE 1.—STANDARDS FOR INTERCONNECTING

DISTRIBUTED RESOURCES WITH ELECTRICAL POWER

Voltage range, 

% nominal

a Maximum clearing time, 

s 

V < 50% 0.16

50% ≤ V < 88% 2.0

110% < V < 120% 1.0

V ≥ 120% 0.16

a Maximum clearing times for DER ≤ 20 kW; Default clearing 

times for DER > 30 kW

Frequency range, 

Hz

Maximum clearing time, 

s 

f > 60.5 0.16

bf < 57.0 0.16

c

59.8 < f < 57.0 Adjustable (0.16 and 300)

b 59.3 Hz if DER ≤ 30 kW

c For DER > 30 kW
NASA/CR—2015-218440 8 

Figure 7.—IEEE 1547 versus NERC PRC-024 (Ref. 9). 

1.2.2 Maritime Power Systems

1.2.2.1 Current Regulations

Modern maritime vessels are advancing power management systems in several areas that are of 

interest to TeDP architecture. Both the shipping industry and TeDP are looking at taking advantages of 

electric propulsion to save energy while investigating different energy storage techniques to power critical 

loads. Some shipping applications resemble a power grid made up of isolated power generators driving 

primarily inductive loads such as the pumping motors on a Floating Production Storage and Offloading 

ship. The shipping industry can provide some insight in isolated microgrids and is already regulated by 

several societies, including the American Bureau of Shipping (ABS), Det Norske Veritas (DNV), and 

Lloyd’s Register. The regulatory bodies have not created standards for a maritime DC power system for 

reasons that will be covered in a later section, though progress is being made in that area.

While MIL-STND-704 guarantees the power quality at the input of utilization equipment, the ABS 

rules for Steel Vessels and for Mobile Offshore Drilling Units (MODU) govern every aspect of the power 

system. The ABS rules govern the response of generators for a given load change with the worst case 

specified or agreed upon by purchaser and manufacturer. As fault scenarios and power flow (load flow) 

solutions depend on the transmission and distribution networks as well as the generators and loads, the 

power system designer must gather and analyze the entire system to ensure that the ABS regulations will 

be met. A significant difference between the military standard and the maritime standards is that the 

shipping standards assume generators share a common bus.

The voltage level on shipping vessels typically varies between 460 V and 15 kV, depending on the 

load application. The lower voltage is used for auxiliary loads such as lighting, communication equipment 

and similar equipment. The higher voltages are used for winches, pumping motors and other large 

inductive loads, connected either directly on-line or through large motor controllers. The ABS specifies 

that the generator voltage is to be within ±2.5 percent of rated voltage for all loads between zero and the 

load at rated power factor. For emergency generators, that limit is increased to ±3.5 percent. Generator 
NASA/CR—2015-218440 9 

transient voltage variations are required to be within the range of –15 to 20 percent of the rated voltage 

given the following load change events:

• A load equal to the starting current of the largest motor or a group of motors, but in any case, at 

least 60 percent of the rated current of the generator, and power factor of 0.4 lagging or less, is 

suddenly thrown on with the generator running at no load 

• A load equal to the above is suddenly thrown off 

The voltage must be restored within ±3 percent of the rated voltage in less than 1.5 s for both load 

acceptance and load shed scenarios. From the above conditions, the ABS considers the maximum load 

change, the load inductance, recovery time and recovery limits during transients when regulating voltage.

The ABS does state what voltage fluctuations will be at the electrical equipment supplied by the 

generators. The electrical loads should operate with a permanent voltage variation of 6 and –10 percent

with a transient variation of ±20 percent with a recovery time of 1.5 s. The difference between allowable 

voltage variations between the generator and distribution system illustrates that the ABS considers how 

much voltage drop and dynamic response the connecting equipment can have the electrical system. The 

voltage continuous variation in DC distribution systems can be ±10 percent. The DC system can have a 

voltage cyclic variation deviation of 5 percent and a voltage ripple of 10 percent. The DC variations 

include systems supplied by rectifiers. As the rectifier output is ±10 percent but the AC generator can be 

–15 to 20 percent, it can be assumed that the ABS expects the rectifier to provide dampening to DC 

system transients. 

The ABS also defines the power sharing between generators since the generators are on a single bus.

The reactive power requirements are of interest as reactive power sharing is typically driven by voltage 

control. The ABS defines the amount of allowable difference in reactive power generation between 

sources and this may need to be considered in the TeDP architecture as the SMES and a generator may 

both inject power into the same bus during a disturbance or fault. 

Energy storage that is used as emergency power on ships is typically provided with batteries. The 

ABS directs that where energy storage is the sole means of supplying DC power equipment for essential 

services, failure of the charging equipment cannot result in the total loss of the energy storage services.

The energy storage should be capable of automatically connecting to the emergency switchboard in the 

event of a failure to the main source of electrical power while immediately supplying power to critical 

services and carrying the emergency electrical load without recharging while maintaining the voltage of 

the energy storage within 12 percent of nominal voltage throughout the discharge cycle. 

The ABS rules also cover frequency, earthing, and protection requirements that may be of interest to 

the TeDP architecture but not directly related to the voltage study. The ABS provides the required voltage 

level for insulation testing on electric machines that are similar to some tests in MIL-HDBK-704. The 

TeDP standard development may want to develop new standards for superconducting machines and 

cables as the superconducting TeDP system will provide novel fault and degradation conditions. 

The Institute of Electrical and Electronic Engineer (IEEE) has developed a recommended practice for 

1 to 35 kV Medium-Voltage DC (MVDC) power systems on ships (IEEE Std 1709) and a recommended 

practice for electrical installations on shipboard (IEEE Std 45). The IEEE states that the common ratings 

of the MVDC power systems, including their operating devices and auxiliary equipment, should be 

selected from the following: 

• Rated maximum voltage 

• Rated withstand voltages 

• Rated continuous current

• Rated short-time withstand current

• Rated duration of short circuit 
NASA/CR—2015-218440 10

The IEEE goes on to state that the voltage should be determined by the desired generator voltage, 

propulsion motor voltage, converter design, load considerations, standard cable ratings, efficiency, and 

arc fault energy. The continuous DC voltage tolerances should be selected considering the normal loads 

and insulation breakdown. The rated withstand voltage is different for the system than for the power 

electronics (the power electronics are detailed in IEEE Std 1662). The choice of withstand voltage allows 

for different voltage performance criteria or overvoltage patters and should be made considering the 

degree of exposure to lightning and switching surge overvoltages, the neutral grounding of the system, 

and the overvoltage limiting devices. Grounding is essential for the MVDC system as the lack of a 

reference point will allow for the presence of leakage currents that may cause an unpredictable DC offset.

IEEE Std 1709 also addresses the stability of the MVDC system by making it clear that the designer 

is required to describe what is meant by stability. To assist, the IEEE defines the following criteria:

1) Time domain criteria exist

a) Transient recovery time

b) Bounded transients (percent of maximum variation) 

c) Absence of limit cycle behavior

2) Frequency domain criteria exist 

a) For example, 6 dB per 30° margins 

b) Frequency domain techniques using a time domain model 

A description of possible stability studies is also provided, some of which have been started by 

developing the dynamic models of the system. In addition to the time-domain analysis of the dynamic 

models, frequency domain analysis and impedance characterization will be crucial in developing a stable 

MVDC power system. The Quality of Service (QoS) is another factor considered in the MVDC power 

system of ships. The QoS is the metric of how reliably the power system provides power to the loads. To 

do this, the loads must be categorized as un-interruptible, short-term interrupt, long-term interrupt, and 

exempt. 

The power quality may be described in part by the voltage waveform. IEEE Std 45 specifies the 

harmonic distortion allowed on the system and on the electric propulsion system, setting limits for both 

total and individual harmonics. IEEE Std 1709 addresses the quality of power on the MVDC bus through 

voltage ripple and voltage tolerance. It sets the limit on the acceptable RMS value of ripple and noise to 

be less than 5 percent per unit. Also, the following parameters should be defined: 

• Maximum non-repetitive peak

• Maximum repetitive peak

• Maximum repetitive peak-to-peak

Finally, it is prudent to note that safety is critical when designing the MVDC power system. There are 

no known international guidelines for safety of operation of MVDC power systems above 3 kV. 

However, IEEE Std 1628 and MIL-HDBK-1025/10 include safety recommendations that can be applied 

to a MVDC system. DC arc fault currents will be significant and models that identify the location and 

severity of arc flash hazard should be developed. The risk of corona discharge is increased with higher 

voltage and higher current levels, and prevention methods will be necessary. The IEEE recommends the 

disconnection and discharge of all power storage devices, in part to reduce the risk of electric shock to 

personnel and equipment.
NASA/CR—2015-218440 11

1.2.3 Current Aircraft Voltage Standards

The standard that establishes and governs the power interface between military aircraft and aircraft 

utilization equipment is MIL-STD-704. The standard regulates voltage levels, frequency, phase, power 

factor, ripple, electrical noise and abnormal conditions for both AC and DC systems that will be available 

at the input terminals of utilization equipment. Electromagnetic Interference (EMI) is not covered by this 

standard and the document does not go into the quality of power generated, transmitted or distributed; it is 

up to the aircraft manufacturer to determine the requirements of the power system to guarantee the power 

levels set forth in MIL-STD-704. MIL-HDBK-704 defines test methods and procedures for determining 

airborne utilization equipment compliance with the electric power characteristics requirements.

AC power is required to be single-phase or three-phase with a wire-connected grounded neutral 

system. While a 400 Hz, fixed-voltage scheme is typical, the standard allows variable frequency and 

double voltage equipment. Variable frequency systems may have frequencies between 360 to 800 Hz with 

a nominal voltage of 115/200 V while double-voltage systems may have nominal voltages of 240/400 V 

with a nominal frequency of 400 Hz. It is noted that the system is not allowed to be variable frequency 

and dual voltage. The phase sequence and markings are specified in the standard as well. The AC system 

is regulated using the following characteristics and are defined for both normal and abnormal conditions: 

• Steady-state voltage

• Voltage unbalance

• Voltage modulation 

• Voltage phase difference

• Distortion factor

• Distortion spectrum

• Crest factor

• DC component 

• Steady-state frequency 

• Frequency modulation 

• Transient peak voltage

• Voltage transient envelope

• Frequency transient envelope 

• Voltage and frequency recovery times

• Power factor

The MIL-STND allows for a two-wire or negative ground return DC system having a nominal voltage 

of 28 or 270 V. The RTAPS TeDP system assumes a three-wire, bipolar system with negative return. The 

following characteristics are defined to govern the operation of a DC system:

• Steady-state voltage

• Distortion factor

• Distortion spectrum

• Ripple amplitude 

• Transient voltage

• Voltage and frequency recovery times

In both AC and DC systems, it is assumed the protection equipment will keep the voltage within the 

operating limits. 
NASA/CR—2015-218440 12

MIL-STND-704 contains several requirements that are paramount to the reliability and safety of an 

airborne electrical system. One requirement is that protective devices operate independently of control 

and regulation. While prudent in a traditional system, this requirement may not be feasible in a DC 

electrical propulsion system as the power electronics in the converters may be used to regulate voltage as 

well as for protection. Indeed a novel DC electrical system may upend the traditional protection scheme, 

removing traditional circuit breakers and relying on fault current limiters and converters for both power 

regulation and protection. The protection and regulation circuits could still use independent 

instrumentation.

The military standard does highlight several areas that should be considered when developing a TeDP 

electrical system. The bonding of all AC devices should be carefully considered to allow for any system 

unbalance. 

1.3 Challenging in Creating DC Propulsion Standards

The marine industry is in the process of exploring DC propulsion systems with several navies funding 

research into the dc power system architecture, protection and energy storage. At this time, these 

regulatory societies have not released standards for a maritime DC power system. The ABS society 

includes some rules on DC propulsion (Ref. 10) and the IEC has several standards (Ref. 11) that include 

requirements for DC systems but there currently is no path in getting a full DC electrical system on a 

maritime vessel approved by any regulatory body The lack of standards has not stopped research into the 

areas and in fact may allow companies to explore the entire design space of DC systems by not 

constraining solutions by over-burdensome regulation. ABB delivered the Dina Star in 2013 that utilizes 

an onboard DC grid that creates a flexible marine power and propulsion system (Ref. 12). Perhaps by 

being the first to market and having the opportunity to prove their design, ABB was granted approval in 

principle for the Onboard DC Grid concept by ABS in January 2014 (Ref. 13). This will allow ABS to 

review the innovative and novel concept of onboard DC systems and provides a path for approval of DC 

systems into the traditional classification rules. 

Outside of commercial applications, the US Navy is trying develop a Medium Voltage DC (MVDC) 

demo that de-risks shared energy storage and advanced MVDC circuit protection, operating above 4 kV 

(Ref. 14). The British Royal Navy has recently published several papers on the challenges in maturing DC 

designs to the point of acceptance in naval vessels (Refs. 15 and 16).

DC systems provide a stability challenge different to AC systems as the presence of reactive power in 

an AC system provides a stabilizing effect. The type of loads are important in a DC systems as constant 

power loads may cause instability in DC systems; as more current is drawn, then voltage has to drop to 

maintain a constant product of voltage and current. The ratio of load and source impedances has to be 

well understood in determining system stability (Ref. 15), a problem bidirectional loads and converters 

exacerbate as the source and loads are interchangeable and the relationship in impedances must also 

reverse which will probably require active control compensations for bi-directional loads. The use of 

capacitance in the network for stability is a common practice but the trade is providing enough 

capacitance to maintain stability while not oversizing capacitors and increasing the cost and size of 

converters. Finally, a low inductance busbar is important in creating an electrically stiff network; possible 

appropriate busbars are in different stages of development. 

Fault protection and clearance is another well-known issue in DC networks. The lack of a zero-crossing 

requires that the switchgear to be large and expensive, especially in converter based systems which can have 

a very high inductance to resistance ratio on short circuit events. Current shipping applications are 

considering using the power electronic devices in the network to provide power regulation and system 

protection. Fold back control strategy allows the source impedance to be dynamically controlled so that it 
NASA/CR—2015-218440 13

follows the load impedance under faulted condition and thereby limiting and eventually eliminating the fault 

current. If the converters are to be used for system protection, they can do so accurately enough to limit the 

currents to a point where traditional over-current and differential circuit breaker discrimination strategies do 

not work making it difficult to isolate the fault (Refs. 16 and 17).

Finally, the role of energy storage on DC maritime systems is not well defined. Commercial 

applications often assume that redundancy in generation will alleviate the need for large energy storage 

systems. Whether the energy storage is located centrally or distributed next to critical loads will impact 

the behavior and control of energy storage and current regulations have taken the approach to define the 

power quality at the critical load’s interface to the power system and have left it up to the designer to 

determine the size, type and control of the uninterruptible power supply based on the duration and duty 

cycle of the load.

1.4 TeDP Electrical System Voltage Standards 

Flight-weight propulsive power rated microgrid systems necessitate the introduction of new aircraft 

distribution system voltage standards. Voltage impacts much of component and system performance. All 

protection, distribution, control, power conversion, generation, energy storage, and cryocooling 

equipment are affected by voltage regulation requirements. Information on the desired operating voltage 

and voltage regulation is required to determine nominal and maximum currents for sizing distribution and 

fault isolation equipment, developing machine topologies and machine controls, and the physical 

attributes of all component shielding and insulation. 

Voltage standards provide assurance that electrical equipment will be operate and integrate 

effectively by applying generally accepted operating limits. These limits impose requirements on system 

components. Existing voltage standards express steady-state and transient limits in order to provide a 

common framework for component manufacturers. Adherence to the standard ensures that the equipment 

will operate effectively within the context of a conventional system. Additionally, the implementation of 

these requirements imposes generally accepted implications on the system. However, the introduction of 

new standards within the context of mature systems is challenging due to implications on the electrical 

systems supply chain. On the other hand, existing standards are insufficient to address the unique needs of 

revolutionary TeDP electrical systems.

The development of a voltage standard requires addressing a system in multiple perspectives. 

Therefore, the standard development activity manages stakeholder expectations and visions and results in 

a common set of bounds which guide the achievement of the goals. These bounds may encompass 

operational envelope, personnel safety, performance objectives, governance authority, or system stability. 

In order to capture expectations and requirements of different stakeholders, it is important to perform 

stakeholder analysis. Figure 8 starts to illustrate the major groups that will have input to the voltage 

standard and what a voltage standard will impact. This study focuses on framing the objectives and 

sensitivities from a design/engineering perspective. 

1.4.1 Regulation, Protection, and Recovery: A Design/Engineering Perspective

Many of the requirements on regulation, protection, and recovery systems are derived from the 

systems transient performance. Therefore, as the challenges associated with custom voltage regulation 

and protection are addressed and matured for TeDP systems, new standards must evolve to meet the 

needs of the industrial base. Fundamentally, for a TeDP system, voltage limits ensure the provision of 

uninterrupted thrust to the aircraft during all flight conditions and operations in a manner which does not 

put operators at risk. For a TeDP system architecture, three areas of voltage management are required. 

These include voltage regulation, fault isolation and protection, and recovery and reconfiguration. 
NASA/CR—2015-218440 14

Figure 8.—Voltage Standard Development Stakeholder Map. 

Recovery solutions were introduced via segregation and symmetry during system architecting to 

ensure that no single-point failure in the TeDP system would lead to a steady state loss in thrust power 

below minimum take-off power requirements. Additionally, the configuration is such that no adverse 

yawing moments are generated during a TeDP system fault.

During normal operation the system is operating with an “absence of any fault or malfunction that 

degrades performance beyond established requirements” (Ref. 18). Under these conditions, power 

conversion equipment manages voltage variations. When a fault or degradation occurs, protection devices 

operate to remove the malfunction within a broader set of abnormal operation limits. A notional voltage 

limit plot is illustrated in Figure 9. 

Manipulation of these normal and abnormal voltage limits determines the requirements for each 

device within the electrical system. Normal limits define the required operation from the power sources 

and conversion/regulation equipment. Abnormal limits determine the impact of the maximum current and 

voltage ratings for all devices, as well as the current interruption capability for protection devices. 

Identifying the preferred voltage range for systems with low TRL components has its own challenges. In 

the absence of data, estimates of component attributes and sensitivity to requirements are based on 

projections of current technologies and first principle estimates regarding future technology capabilities. 

1.4.2 Operational Voltage Limits Category Definitions 

Considering the unique attributes of the loads and redundancy available on a TeDP electrical system, 

the definitions typically applied for aircraft power systems deserve revisiting. The definitions presented 

here are augmented from MIL STD 704F (Ref. 19).
NASA/CR—2015-218440 15

Figure 9.—Steady-state and Transient Voltage Limits for Normal and Abnormal Operations.

1.4.2.1 Normal Operation

Normal operation occurs when the system is operating as intended in the absence of any fault or 

malfunction that degrades performance beyond established requirements. 

The major adaptation to this definition is the removal of transfer operation as a subset of normal 

operations. As is illustrated in Figure 9, voltage variability for normal operations can be very narrow. 

Nominal loads on a TeDP system are highly regulated and coordinated. Other types of microgrid 

networks must interface with loads which are unscheduled and unpredictable. Under normal conditions, 

the only transients experienced by the system are from the acceleration and deceleration of propulsor fans. 

Even in the presence dynamic loading requirements from stability augmentation for flight controls or 

from torque loads imposed by inlet distortions, the transients are relatively slow and can be coordinated at 

the system level.

1.4.2.2 Abnormal Operation

Under fault conductions a subset of the propulsors shall be permitted a degradation or loss of 

function as specified by aircraft propulsion and control requirements. Electrical system equipment shall 

not suffer damage or cause an unsafe condition. 

Limits on the abnormal operations enforce the magnitude of the disturbance allowable. When normal 

operation limits are exceeded, protection and controls equipment act to isolate the failure and return to a 

normal operating state by initiating recovery and transfer operations. 

0.001 0.01 0.1 1 10 100

Voltage (Normalized)

Time from Onset of Transient (Normal) or Over/Under Voltage (Abnormal) 

[seconds]

 

Normal Operation Upper Limit

Normal Operation Lower Limit

Abnormal Operation Upper Limit

Abnormal Operation Lower Limit

Regulation

1 Nominal

Protection/Recovery Transient Steady-State
NASA/CR—2015-218440 16

1.4.2.3 Recovery Operation

After experiencing and abnormal operation, equipment undergoes recovery. Requisite propulsors 

shall automatically resume specified performance when normal operating characteristics are resumed. 

The requisite propulsors are determined by airframe by propulsion and control requirements. 

Recovery operations occur on the faulted branches of the TeDP system after an abnormal operation 

occurs. Temporary loss of power to the propulsors will result during fault isolation and potential 

deployment of uninterrupted power supply (UPS) energy storage (depending on the location of the fault 

and fault zonal protection). After fault clearance, attempts may be made to re-engage the previously fault 

equipment. However, with fail-safe redundancy requirements for OEI scenarios, faulted sources may be 

forfeit for the remainder of the aircraft mission after a fault. Power for the propulsors allocated originally 

to the faulted source branch can receive power from healthy branches via transfer operations. 

Following isolation and clearance of a load side fault, recovery may then include the removal of 

thrust requirements from propulsors with faulted electrical equipment or damaged mechanical equipment. 

1.4.2.4 Transfer Operation

Normally operating equipment undergoes a transfer operation following abnormal operation of other 

equipment to facilitate a recovery operation.

Recovery operations occur on the faulted branches of the TeDP system. This operation occurs when 

power is routed from a healthy branch in response to an abnormal event on an adjacent branch. A transfer 

operation introduces the largest non-fault-related step loads on the systems. However, the timing of these 

loads may be coordinated with energy storage charge/discharge operation.

1.4.3 Selection of the Optimal Operating Voltage

With the bulk of the system components being superconducting, the requirements for an airborne 

TeDP system may not be driven by interface to normally conductive systems. Therefore, additional 

freedom is available to more intelligently define operational attributes of this wholly superconducting 

system. In the absence of applicable design standards, this requires a fundamentally different approach in 

defining optimal configuration. Due to the integrated nature of this system, the definition of the optimal 

voltage standards must be performed holistically. The sensitivities of all system components must be 

considered simultaneously. 

Figure 10 illustrates the process and framework developed for identifying the optimal operating 

voltage standards for non-conventional aerospace microgrid systems. This process and framework is 

applied to the Rolls-Royce N3-X TeDP architectures to identify the trend for the optimal voltage levels 

based on predictions of component voltage mass and efficiency sensitivity, integrated system-level effects 

of voltage levels, and future technology development factors. With this information in hand, more 

accurate assessment of electrical system weight, volume, and reliability may be generated and drive 

component development requirements. The scope of the 2012 RTAPS study (RTAPS TEDP I) and the 

scope of the current RTAPS study (RTAPS TEDP II) are indicated in this figure. 

This process for selecting the preferred operating voltage follows the traditional top-down, bottom-up 

engineering V-process (Ref. 20). This process is defined in four phases: architecture definition, nominal 

voltage selection, transient limit selection, and recovery scenario analysis. The tasks and activities for 

each of these phases. Each phase of this process requires additional levels of fidelity to evaluate the 

impact of voltage limits on the architecture weight and efficiency. The method and results discussed in 

this paper focus on the second phase of this voltage standard definition process. The architecture 

definition portion of this process has been documented in Reference 21.
NASA/CR—2015-218440 17

Figure 10.—TeDP Architecture Design Approach. 

1.4.3.1 Phase 1

Phase 1 of this process (Architecture Definition) focuses primarily on defining the context and 

content to be considered when defining the transient and steady state voltage limits. This requires that the 

architecture be defined in terms of its general structure, functional interfaces, and technologies. 

Conceptual level modeling of power flow and sizing are used to justify the concept selection. The sizing 

of these systems requires that general contingency response concepts be defined for all pertinent operating 

scenarios. 

The Phase I tasks were performed for this the N3-X architecture during previous studies. In 2012, the 

Rolls-Royce Electrical Power and Control Systems (EPACS) group delivered a Research and Technology 

for Aerospace Propulsion Systems (RTAPS) study to NASA that considered the stability, transient 

response, control, and safety of a high-power electric grid for the NASA N3-X TeDP system (Ref. 22). 

Under this research contract, promising electrical system architecture concepts were identified and 

compared. Weight and complexity comparisons were made between systems concepts based on estimated 

future component weight trends (Refs. 23 and 24). 

This effort was successful in detailing the impact of safety, reliability, redundancy, protection, and 

integrated flight control requirements on TeDP system design. However, the holistic perspective applied 

by Rolls-Royce in this study identified significant gaps that must be addressed for this nonconventional 

electrical system development. The operating voltage level for this aircraft was discussed identified as an 

area interest for further evaluation. 

1.4.3.2 Phase 2

Identifying the optimal voltage for the N3-X TeDP architecture is the primary objective of this study. 

During Phase 2 (Steady-State Voltage Selection and Sizing), additional scrutiny is applied to the 

architecture selected in Phase 1. The Phase 2 tasks are: component decomposition, subcomponent 

sensitivity assessment, system sizing and steady-state voltage selection. The activities in this phase map 

the effect of system level operating parameters on the individual subcomponents of the system. The 

sensitivities of these subcomponents are then evaluated against variations in the operating parameters. 
NASA/CR—2015-218440 18

These sensitivities are expressed in the form of first principle subcomponent models of or data where 

available. Once the subcomponent sensitivities are defined, an integrated system model that assembles 

these components for overall system evaluation is constructed. This evaluation determines the expected 

mass, efficiency, and operating parameters as a function of voltage and other system level operating 

parameters (temperatures, shaft speeds, etc.). This phase applies assumptions regarding the impact of 

protection and recovery on component size

One major challenge in identifying the optimal operating voltage will be the identification of 

component and system sensitivity to voltage regulation parameters. The process for performing this 

analysis will follow the basic framework for selecting the optimal dc system voltage used by Christou 

et al. and Cotton et al. (Refs. 25 and 26). Within the constraints of a fixed duct area size, minimum 

clearance required between wires and ground, and standard wire gauge sizes and their associated 

diameters, ac and dc system current rating, and insulation thickness, cases were considered for 

distribution systems for the configurations. The results of these system options was compared for peak 

voltage rating versus conductor diameter, power rating versus single-wire voltage rating, and power-toweight ratio versus single-wire peak voltage rating. The wire voltage rating varies nonlinearly with power 

rating and power-to-weight ratio. Such trends allow the system to be optimized for the operating voltage 

based on maximum power rating or power-to-weight ratio. Alternatively, an operating voltage can be 

selected that is a compromise between these and other identified objectives. 

A similar approach to voltage selection was applied in this study. However, the principles of this 

study are expanded to consider each piece of equipment in the electrical system. Therefore, reasonable 

estimates of the effect of voltage on power densities for superconducting generators and motors, cable, 

fault-current limiters, circuit breakers, and cryogenic converters are required. Where available, 

cryogenically operating component data was used for sensitivity models (IGBTs, diodes, superconducting 

cables). However, in most cases parametric models based on first principle and published analytical 

estimates were developed to identify the mass, efficiency, and voltage sensitivity of system components.

1.4.3.3 Phase 3

Many of the requirements on protection and recovery systems are derived by the transient 

performance of the system. Therefore, Phases 3 and 4 require dynamic system models and simulations. 

The activities in Phase 3 (Protection Configuration and Transient Limit Selection) focus on refining the 

requirements on the protection devices. The magnitude of the transient overcurrent and overvoltage 

requirements are assessed with respect to the performance of the protection devices (e.g., interruption 

time for solid-state circuit breakers (SSCBs) and resistivity transition for SFCLs). At the conclusion of 

this phase, the sizing sensitivity is revisited with updated transient protection equipment requirements. 

Parallel to the voltage sensitivity modeling activities in support of Phase 2 objectives, dynamic 

models were developed for all grid components to enable trade studies around voltage regulation, fault 

protection and isolation, and thrust power recovery. The development of these models is presented in this 

report. However, addition work is necessary to exercise the models against fault conditions to update 

protection and conversion component sizing requirements. 

1.4.3.4 Phase 4

The final phase of this process (Recovery Verification and Refinement) is intended to confirm and 

refine the contingency strategies defined in Phase 1. Assuming that the TeDP system must operate as an 

UPS, dynamic simulations of system recovery are performed to update energy storage and generation 

requirements for these recovery scenarios. Upon completion of this analysis, the sizing and sensitivity is 

performed with these additional requirements updates. 
NASA/CR—2015-218440 19

1.5 Introduction Summary

Establishing the feasibility and viability of a DC, superconducting, DC, microgrid TeDP system 

requires detailed understanding the integrated performance of electrical components, as well as the 

implications of system level requirements. In the absence of refined standards regarding voltage 

regulation, fault protection and isolation, and system recovery for this revolutionary system concept, one 

must rely on model representations of the system to guide technology development. This study supports a 

holistic TeDP electrical system architecture design approach through the development of parametric and 

dynamic models for the entire N3-X TeDP system. The parametric sizing models are used to determine 

the system level impact of performance requirements in terms of overall system mass and efficiency. 

Following preliminary sizing, the dynamic models are used to determine the transient operation 

requirements for the equipment. This information refines the assumptions used during parametric sizing 

trades to refine the architecture performance estimates. 

This document reviews the development of these models and their implementation for nominal 

operating voltage optimization and dynamic analysis of fault accommodation strategies. It is hoped that 

this process and continued development of these tools will assist in establishing limits on normal and 

abnormal electrical system operations. This information will provide useful in providing requirements for 

individual technology development and provide continued support for the definition of TeDP system 

voltage standards. 
NASA/CR—2015-218440 20

2.0 Architecture Selection

2.1 Architecture Candidates

Candidate architecture concepts were adapted from deliverables provided under the previous RTAPS 

task order. These architectures are described. All architecture concepts here were sized considering failsafe and single point failure requirements. No single point failure will lead to, or yield a catastrophic loss 

in thrust. 

2.1.1 Concept 1: Baseline Architecture

A baseline architecture concept is illustrated in Figure 11. The baseline consists of four independent 

electrical systems. Each electrical system consists of 1 generator, 1 AC/DC converter, 1 bus with an 

associated energy storage device, and 4 propulsors. The initial concept included a total of 14 propulsors 

which required two buses supporting 3 propulsors and two buses supporting 4 propulsors. In order to 

mitigate asymmetric thrust with bus and generator faults, an even number of propulsors is desirable for 

each bus. Therefore, the number of propulsors was increased to 16, with four assigned to each power bus. 

This alteration is consistent for all architecture concepts. 

The engine out scenario produces the sizing case for this architecture. In this scenario a group of 8 

propulsors must provide 100 percent of the required thrust, supported by 2 buses. Therefore, each 

propulsor much account for roughly 12.5 percent of the minimum power and each bus must be able to 

support 50 percent of the power. 

Figure 11.—Baseline Architecture Diagram. 
NASA/CR—2015-218440 21

This simple radial architecture concept acts as the baseline architecture for architecture comparisons. 

While this architecture is attractive in terms of simplicity, it is evident that the components are oversized.

2.1.2 Concept 2: Inner Bus Tie Concept

The second architecture concept allows for a reduction in propulsor oversizing by including a single 

point of reconfigurability in the system. 

While the engine out scenario remains sizing critical, closing the bus tie engages more propulsors to 

provide thrust. Each set of 4 propulsors must provide a 1/3 of the total propulsive power. Therefore, the 

inner and outer buses and generators are sized differently to reflect the required interconnectivity 

(Figure 12).

2.1.3 Concept 3: 3-Bus Multifeeder Concept

A further reduction in propulsor oversizing can be achieved by allowing power to be rerouted through 

secondary feeders from other alternative bus sources. The engine out scenario no longer produces sizing 

critical requirements for the propulsor system. For this concept, the propulsors are sized to mitigate two 

propulsor fault conditions (Figure 13).

2.1.4 Concept 4: Cross-Redundant Multifeeder Concept

The final concept evaluated under the previous task order includes multiple layers of interconnectivity 

to reduce distribution systems oversizing. This comes at the cost to complexity and increases to the 

number of components in the system. 

Figure 12.—Inner Bus Tie Architecture Diagram. 
NASA/CR—2015-218440 22

Figure 13.—Three-Bus Multifeeder Architecture Diagram. 

This concept benefits from having an equal number of propulsors for each bus (Figure 14 and 

Figure 15). This allows for consistent sizing of all primary and secondary feeders, as well as the 

transmission and distribution components. 

2.1.5 Concept 5: 4-Bus Inner Bus Tie Multifeeder Concept

An additional concept architecture was generated by combining the reconfigurability employed by the 

inner bus tie and multifeeder concepts. 

This architecture was introduced because of its ability to represent three or the four candidate 

architecture in its modeling. The baseline architecture is represented by disconnecting the secondary 

feeders and opening the bus tie. Additionally, the inner bus tie concept is represented by disconnecting the 

secondary feeders. Finally, the multifeeder concept is represented by closing the bus tie and connecting 

the secondary feeders. Scaling of component capability limits is also required to represent these concepts. 
NASA/CR—2015-218440 23

Figure 14.—Cross-Redundant Multifeeder. 

Figure 15.—Four-Bus Inner Bus Tie Multifeeder. 
NASA/CR—2015-218440 24

2.2 Architecture Evaluation and Selection

Table 2 shows the evaluation criteria used for evaluation and selection of the architecture concepts. 

The criteria in yellow were used in comparing and selecting the architecture of interest during the 

previous RTAPS task order. These criteria consider the architectures weight, complexity and 

performance. The weight approximations were made in the previous study on a specific weight basis and 

without consideration to voltage sensitivities. The weights and complexities of the various architectures 

will change on the voltage range for the system. As such, the analysis performed on the selected 

architecture evaluation remains valid for the other candidates. This evaluation has been performed 

without consideration of voltage sensitivity.

In addition, the current task order required a broadened evaluation of the architectures. With added 

deliverables to develop a process for architecture evaluation, the criteria in green were introduced. These 

new evaluation criteria consider the adaptability/scalability of a process developed by considering this 

single architecture concept (Table 3). Additionally, the difficulty involved in modeling the architecture 

was also considered.

Weightings were applied to the evaluation criteria to determine the preferential architecture going 

forward in for this project. A 1, 3, 9 scaling was applied for low, medium, and high weightings, 

respectively. 

Application of these weighting and architecture assessments are illustrated in Table 3. The colors in 

this table indicate the criteria based assessment relative to the baseline concept. The darker the red the 

cell, the worse the concept performs relative to the baseline. The darker the green, the better the concept 

performs. 

TABLE 2.—ARCHITECTURE COMPARISON MATRIX

Architecture Evaluation Criteria

Weighting

Baseline Inner Bus

Tie

3‐bus

Multi‐

feeder

4‐bus

Inner Bus

Tie, Multi‐

feeder

Cross‐

redundant

multi‐

feeder

Previous

Metrics

Weight, kg High 5086 4227 4144 4176 4010

Complexity

Failure Response (rerouting complexity) Low + + ‐

Component count High 116 118 158 160 174

Excess Power, hp Med 100% 33.33% 14.28% 14.28% 14.28%

New

Metrics

Scalability of Process

Breadth of Protection Scheme Med 0 + + ++ +++

Switching Functionality represented Med 0 + + + +

Load interruption represented High 0 0 0 0 0

ModelComplexity

Dynamic model (with protection) High 0 ‐ ‐ ‐ ‐‐‐

Multiple sources on common bus Med 0 + ++ ++ +
NASA/CR—2015-218440 25

TABLE 3.—ASSESSMENT MATRIX

The weighted sum for each concept and the weighted sum relative to the baseline are given below in 

the assessment matrix.

Two architecture concepts are evaluated to be preferential to the baseline architecture and two 

concepts are evaluated to be worse than the baseline (Figure 16). 

The 4-bus, inner bus tie, multifeeder concept is narrowly the preferential architecture according to 

this assessment. This is particularly due to the breadth of the protection schemes that it represents (the 

baseline, inner bus tie, and 3 bus multifeeder concepts can all be represented with the same model 

structure as this concept). Additionally, for some protection schemes and failure scenarios, this 

architecture requires that a load receive power from both the primary and secondary feeders 

simultaneously. The 4-bus, inner bus tie, multifeeder architecture also exhibits comparable weight and 

complexity scores to the preferential architecture which was selected previously (inner bus tie concept). 

2.3 Weight Sensitivity and Deliverable Objectives

The design of superconducting transmission equipment current is predominately driven by objective 

to minimize the cost and volume of distribution cables with increased efficiency. However, for an 

airborne superconducting microgrid, overall system weight becomes more critical. 

Considering the weight breakdown for this TeDP microgrid performed in the previous Rolls-Royce 

RTAPS study (Ref. 27) with the indicated assumptions of power, current, and torque density in Figure 17. 

For these architectures, cable weight is not of primary concern due to their relatively small overall weight 

contribution (4.1 percent of the system weight on average for all architectures considered). Looking at 

these weight evaluations, the voltage sensitivity of power electronics is preeminent (Table 4). This is 

followed by machine and protection sensitivity. Understanding the impact of voltage on these components 

is required to determine the optimal operating voltage for this TeDP architecture. 

This deliverable does not explore these voltage sensitivities. However, it reviews typical and 

projected voltage levels demonstrated and studied for these architecture components. Additionally, the 

deliverable also presents an architecture down selection for our future sensitivity, sizing, and dynamic 

modeling. 
NASA/CR—2015-218440 26

Figure 16.—Relative Weighting of the Architectures. 

30,000 hp Min Power Required 40 kW/kg Power Electronics Power Density 

4500 rpm Prop. Speed 100 Nm/kg Motor/Generator Torque Density 

7500 rpm Turbine Speed 200 kW/kg DC Breaker Power Density 

±10 kV Voltage 350 kW/kg AC Breaker Power Density 

500 A/kg/m Feeder Linear Current Density 0 lb/ft Yaw Trimming Moment

Figure 17.—Weight Breakdowns. 

TABLE 4.—ARCHITECTURE WEIGHT BREAKDOWN 

[Results do not include energy storage weight estimates or fault current limiter weight.]

Arch1: 

Baseline, 

%

Arch2: Inner 

bus tie, 

% 

Arch3: 3-bus 

multifeeder, 

% 

Arch4: 

Crossredundant 

multifeeder,

%

Arch5: 4-bus 

inner bus tie, 

multifeeder, 

% 

Arch5: 4-bus 

inner bus tie, 

multifeeder,

% 

Average,

% 

Generators 11.2 13.5 13.7 14.2 13.6 13.6 13.3

Converters 22.0 26.5 27.0 27.9 26.8 26.8 26.2

Distribution system 3.5 3.9 4.3 4.3 4.3 4.3 4.1

Motor drives 22.0 17.6 15.4 15.9 15.3 15.3 16.9

Motors 18.7 15.0 13.1 13.5 13.0 13.0 14.4

Protection equipment 20.1 19.7 23.4 20.9 23.2 23.2 21.7
NASA/CR—2015-218440 27

2.4 Overview and Naming Convention for Selected TeDP Architecture

The four-bus, inner bus tie, multifeeder architecture employs multiple layers of redundancy to provide 

uninterrupted thrust power the aircraft during electrical system failures. Source redundancy is provided by 

two engines, each driving redundant electric machines. Each engine is sized to provide the overall 

minimum thrust power required. Therefore, each electric machine is sized to provide half of the required 

thrust power. Under this arrangement, the transients experienced by the gas turbine will be a sudden 

increase in torque from 50 to 100 percent in the event of a one engine inoperable (OEI) condition or a 

torque loss from 50 to 25 percent for a single point electrical system failure. 

During nominal operation, each generator supports a single distribution bus. Each radial connection 

between the generator source and the bus is termed a branch. Each branch is sized to generate and 

distribute a quarter of the required thrust power nominal and half of the thrust power required during a 

faulted condition. Power is delivered to the propulsors by means of primary and secondary feeders. 

Secondary feeders are used during branch and OEI fault scenarios. Additionally, the bus tie allows a 

single engine to provide power to all of the propulsors. Further discussion on the rerouting of power in 

response to failures is provided in Section 7.6. 

Figure 18 shows a more detailed diagram of the architecture selected for this study. This figure also 

indicates the naming convention for each of the components in the system. 

Figure 18.—4-Bus, Inner Bus Tie, Multifeeder Architecture Diagram with Component Naming Convention.

G-L1 G-L2 G-R2 G-R1

M-L1 M-R1 M-L8 M-R8 M-L3 M-R3 M-L6 M-R6 M-L4 M-R4 M-L5 M-R5 M-L2 M-R2 M-L7 M-R7

Left 

Engine

Right 

Engine

Left Side Superconducting 

Propulsor Motor

Right Side Superconducting 

Propulsor Motor

GC-L1 GC-L2

B-L1 B-L2

ES-L1 ES-L2

PF-L1-L1

PF-L1-R1

PF-L1-L8

PF-L1-R8

PF-L2-L3

PF-L2-R3

PF-L2-L6

PF-L2-R6

PF-R2-L4

PF-R2-R4

PF-R2-L5

PF-R2-R5

PF-R1-L2

PF-R1-R2

PF-R1-L7

PF-R1-R7

SF-L1-L3

SF-L1-R3

SF-L1-L6

SF-L1-R6

SF-L2-L1

SF-L2-R1

SF-L2-L8

SF-L2-R8

SF-R2-L2

SF-R2-R2

SF-R2-L7

SF-R2-R7

SF-R1-L4

SF-R1-R4

SF-R1-L5

SF-R1-R5

Propulsor Side 

Feeder Breakers

(CB-PF- P_ _, 

CB-SF-P_ _)

Propulsor Side AC 

Breakers

(CB-AC-P_ _)

CB-AC-GL1

SFCL-AC-GL1 

CB-GS-TL1

PMC-L1PMC-R1PMC-L8PMC-R8PMC-L3PMC-R3PMC-L6PMC-R6PMC-L4PMC-R4PMC-L5PMC-R5PMC-L2PMC-R2PMC-L7PMC-R7

Propulsor Side 

Feeder Bus

(B-P_ _)

ES-R2 ES-R1

CB-GS-TL2 CB-GS-TR2 CB-GS-TR1

SFCL-AC-GL2 SFCL-AC-GR2 SFCL-AC-GR1 

CB-AC-GL2 CB-AC-GR2 CB-AC-GR1

CB-BS-TL1 CB-BS-TL1 CB-BS-TR2

GC-R2 GC-R1

SFCL-TL1 SFCL-TL2 SFCL-TR2 SFCL-TR1 

CB-ES-L1 CB-ES-L2 CB-ES-R2 B-R2 B-R1 Bus Side 

Feeder Breakers

(CB-PF-_ _-__ ,

CB-SF-_ _-__ )

Normally Closed SSCB

Normally Open SSCB

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL SFCL

SFCL SFCL SFCL SFCL SFCL SFCL

CB-BS-TR1 CB-ES-R1

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

SFCL

Propulsor Side AC SFCL

SFCL’s

(SFCL-AC-P_ _)

SFCL

SFCL

Feeder SFCL

(SFCL-PF-_ _-__ ,

SFCL-SF-_ _-__ )

Wholly Superconducting SFCL

Generator Rectifier

Inverter Superconducting 

Magnetic Energy Storage

Superconducting 

Fault Current Limiter

SFCL-BT-L2-R2 

CB-BT-L2-R2 
NASA/CR—2015-218440 28

Not all components included in this diagram will be necessary in the final architecture. A protection 

system for this architecture may be provided by a combination of circuit breakers, fault current limiters, 

and the converter switches. However, following a dynamic evaluation of system faults, the final 

architecture may only require a subset of the protection devices illustrated in this diagram. Further 

discussion on potential configuration of the protection is provided in Section 6.3. 

The naming convention illustrated in Figure 18 assigns nomenclature based on the relationship of the 

device to its related branch and/or proposal motor. The branches are labeled based on which 

turbogenerators provide them power. This nomenclature is given in Table 5. 

TABLE 5.—NAMING CONVENTION FOR TEDP ARCHITECTURE COMPONENTS

Device Label

Generator G - <Branch Index>

Rectifier GC - <Branch Index>

Bus B - <Branch Index>

Transmission Line T - <Branch Index>

Energy Storage ES - <Branch Index>

Primary Feeder PF - <Branch Index> - <Propulsor Index>

Secondary Feeder SF - <Branch Index> - < Propulsor Index>

Propulsor Bus B - P<Propulsor Index>

Inverter PMC - <Propulsor Index>

Propulsor Motor M - <Propulsor Index>

Turbogenerator AC Circuit Breaker CB - AC - G<Branch Index>

Source Side Transmission Line Circuit Breaker CB - GS - T<Branch Index>

Bus Side Transmission Line Circuit Breaker CB - BS - T<Branch Index>

Primary Feeder Bus Side Circuit Breaker CB - PF - <Branch Index> - <Propulsor Index>

Secondary Feeder Bus Side Circuit Breaker CB - SF - <Branch Index> - <Propulsor Index>

Primary Feeder Propulsor Side Circuit Breaker CB - PF - P<Propulsor Index>

Secondary Feeder Propulsor Side Circuit Breaker CB - SF - P<Propulsor Index>

Propulsor AC Circuit Breaker CB - AC - P<Propulsor Index>

Energy Storage Circuit Breaker CB - ES - <Branch Index>

Bus Tie Circuit Breaker CB - BT - <Branch Index1> - <Branch Index2>

Turbogenerator AC Superconducting Fault Current Limiter SFCL - AC - G<Branch Index>

Transmission Line Superconducting Fault Current Limiter SFCL - T<Branch Index>

Primary Feeder Superconducting Fault Current Limiter SFCL - PF - <Branch Index> - <Propulsor Index>

Secondary Feeder Superconducting Fault Current Limiter SFCL - SF - <Branch Index> - <Propulsor Index>

Propulsor AC Superconducting Fault Current Limiter SFCL - AC - P<Propulsor Index>

Bus Tie Superconducting Fault Current Limiter SFCL - BT - <Branch Index1> - <Branch Index2>
NASA/CR—2015-218440 29

3.0 Terrestrial Systems Benchmarking

3.1 Power Transmission Superconducting Cable Installations

The voltage, current, and power ratings for current ground based power transmission installations 

using superconducting cabling were reviewed. The primary sources for these overviews were Electric 

Power Research Institute’s (EPRI) “Superconducting Power Equipment Technology Watch 2012,” 

(Ref. 28) and two reports from Sumitomo Electric Industries (SEI): “Present Status of International 

Standardization Activities for Superconductivity” (Ref. 29) and “Present Status and Future Perspective of 

High-Temperature Superconductors” (Ref. 30). The information from these reviews is plotted in the 

Figure 19, Figure 20, and Figure 21. The rated current for all of these systems ranges from 0.8 to 10 kA 

and the operating voltage ranges from 1.3 to 275 kV. 

Figure 19.—Voltage, Current, and Power for Existing Terrestrial Cable Installations. 

Figure 20.—Power versus Voltage Trends for Existing Terrestrial Installations. 

1

10

100

1000

1 10 100 1000

Power (MW)

Voltage (kV) 

 

AC Installations

DC Installations
NASA/CR—2015-218440 30

Figure 21.—Narrowed Range for Power versus Voltage Trends for Existing Terrestrial Installations. 

The majority of superconducting installations interface to the electrical grid and are configured to 

operate with three-phase alternating current. These installations are labeled in orange in Figure 21. There 

are several DC transmission systems installed in Russia, China, Japan, and South Korea which are labeled 

indicated in green.

All of the transmission systems compared in these figures utilize LN2 cooled YBCO or BSCCO with 

one exception. This exception is Russia’s Hybrid Energy Transfer Line (ETL), which consists of a 12 m 

MgB2 system cooled by LH2. 

From Figure 22 it is clear that the sizing of the superconducting transmission systems have a definite 

correlation with the rated power of the system. Generally, higher power systems implement higher 

voltage levels. However, focusing on the power levels applicable to TeDP (Figure 21), this trend is not as 

clear as voltages typically range between 1.3 and 35 kV. While this range will act as a baseline for future 

work in this study, all observed trends for the systems considered in this section are exclusively 

applicable to terrestrial systems. In contrast, weight metrics are of more significant interest for an airborne 

TeDP electric grid system.

3.2 Future DC Power Transmission Installations 

In addition to its ability to provide a bulk power transfer with a significantly smaller footprint, 

superconducting DC power transmission systems promise to improve safety, reliability, and efficiency 

relative to existing AC power grid (Ref. 31). In advance of potential future applications, major 

superconducting cable manufacturers are beginning to enhance their high voltage DC capabilities. 

Current work toward Korea’s JeJu Island’s ±80 kVdc substation interconnect and future concept 

plans toward New Mexico’s Tres Amigas 200 kVdc energy hub are two examples of the future 

superconducting state of the art. While the risk and cost associated with current superconducting 

protection and conversion technology prohibited the implementation of a HTS solution for the Tres 

Amigas Project (Ref. 32), superconducting technology advances can have dramatic impacts on terrestrial 

transmission systems in the very near future. 

In their report on Superconducting DC Cable, EPRI describes there baseline future concept for power 

transmission. This concept is described as “an interregional, superconducting dc cable system that is 

intended to achieve 10 GW power capacity with a nominal current and voltage of 100 kA and 100 kV” 

(Ref. 33). While the voltage level is certainly achievable with today’s technology, technology 
NASA/CR—2015-218440 31

improvements are required to allow for increasing the standard operating currents by an order of 

magnitude. According to EPRI: 

“The insulation level of 100 kA is easily achieved with currently available insulation schemes. 

In fact, the voltage level is so low that insulation thickness is determined by structural capabilities 

and ruggedness rather than by voltage standoff capabilities. A higher voltage could be readily 

achieved and would help meet the challenges posed by the high current. However, the advantages 

of keeping the voltage as low as possible are not to be ignored... High-power transmission at 

relatively low voltage is a hallmark of superconducting power transmission systems, both ac and 

dc, and is a key component of their economic viability.” (Ref. 33)

While the superconducting cable of the future requires multiple layers of conductor, insulator, 

formers, and coolant passages, the largest contributor to cable weight and size is the quench conductor. 

Additionally, the cable size is just a small portion of the overall vacuum sealed conduit used to thermally 

isolate the cryogenic system (Ref. 34).

With limited benefits achieved with increased voltage in terms of conductor and insulation impact, 

the main determining factor for voltage decisions is managing quench conditions. 

A lower end limit of the operating voltage may be defined considering the power level of the TeDP 

electric grid and the maximum operating current. The maximum conventional terrestrial distribution 

system current identified was for a Chinese alumina electrolyzer plant. This system’s operating current is 

greater than 10 kA. A cable transmission system providing 25 MW of power with a maximum current of 

10 kA would require a minimum operating voltage of 2.5 kV (or ±1.25 kV). 

Applying EPRI’s future superconducting power transmission target of 100 kA, the operating voltage 

could drop to as low as 250 V (or ±125 V).

3.3 Superconducting Fault-Current Limiter Installations and Prototypes 

There are many installations of transmission and distribution voltage-level superconducting faultcurrent limiters (SFCLs). In addition to these high power installations, there are a few lower power research 

projects involving the design and development of SFCLs at the University of Manchester, a Rolls-Royce 

University Technology Center (UTC). Key facts about each project are listed in Table 2. The Ph.D. 

dissertation from a student at the University of Manchester describes in detail the design, development, and 

test of a SFCL with an integrated vacuum interrupter (Ref. 35). This thesis also describes and tabulates the 

data in Table 2. The references for those sources are individually cited in this report. 

From Table 2, it is notable that the projects using the superconductor MgB2 operate at much lower 

voltage, current, and power levels than the installations using BSCCO and YBCO. This is due in part to 

the more recent discovery of MgB2 as a superconductor (2001 compared to mid-1980s for BSCCO and 

YBCO) and the need for development to scale its production for higher current and power applications. 

All of these applications are for AC terrestrial grid fault-current limiting where the SFCL is the only 

dedicated superconducting device and interfaces to a normally conducting distribution line. 

3.4 High Power Normally Conducting Solid-State Switchgear

While solid-state circuit breakers and switch gear for superconducting or cryogenic systems are only 

in the early research stage, there are initiatives to drive the design and development of normally 

conducting hybrid solid-state circuit breakers for power distribution voltages. Some advantages of high 

power solid-state circuit breakers over mechanical circuit breakers that also apply to aircraft power 

systems are the potential to eliminate momentary interruptions, provide instantaneous current limiting, 
NASA/CR—2015-218440 32

clear faults more quickly, and limit inrush currents for capacitive loads. In Reference 36, EPRI outlines 

requirements and specifications for a 15 kV, 600 A steady-state current rating design. Other relevant 

requirements are a short-circuit current rating of 12.5 kA (symmetrical, for 1 s), less than 10 kW of losses, 

and fault clearing within half a cycle (8.3 ms). 

3.5 Cryogenic Semiconductors

Semiconductor performance is directly related to temperature across several aspects. Mainly, carrier 

density decreases, carrier lifetime decreases, and carrier mobility increases with decreasing temperature. 

First, a doped semiconductor typically decreases in resistance with decreasing temperature due to the 

reduction of lattice scattering of carriers (Ref. 37). Yet, at a specific dopant concentration and material 

temperature, impurity scattering causes a decrease in carrier mobility, increasing resistance, with a 

decrease in temperature. This change in resistance gradient with respect to temperature occurs at 

approximately 50 K for silicon as seen by the relative invariance of electron mobility in the 30 to 50 K 

range in Figure 22 and several orders of magnitude decrease in dopant concentration in the 30 to 50 K 

range in Figure 23. 

Second, a PN junction voltage tends to increase with decreasing temperature due to the increased 

band gap energy (Refs. 37). Furthermore, the saturation current exponentially decreases with decreasing 

temperature. 

The solid state device overall performance at cryogenic temperatures is a complex combination of the 

carrier concentration, carrier mobility, band gap energy, and the device structure. For instance, as the 

temperature decreases from room temperature to 70 K, the threshold voltage of non-punch through 

insulated gate bipolar transistors (NPT-IGBT) increases and the “on” resistance decreases due to the 

increase in band gap and the increase in carrier mobility, respectively. When cooled further, there is a two 

stage resistivity and increasing threshold voltage. Yet in general several groups of scientists and engineers 

have shown that with decreasing temperature, device resistance decreases, device switching speed 

increases and device threshold voltage increases (Refs. 38, 39, and 40).

Figure 22.—Electron Mobility versus Temperature for Different Doping Levels.
NASA/CR—2015-218440 33

Figure 23.—Various Estimations of Ionized Donor Concentration as a Function 

of Temperature. The Background Doping Density Is Assumed To Be 1014

cm-3 and Phosphorous as the Dopant Species.

From literature review, as a generalization for all devices, switching time, thus switching losses, 

decrease by tenfold and the hold-off or reverse break down voltage decreased by ~30 percent, while the 

resistance and threshold voltage combined to give negligible change in conduction losses. The reduction 

in switching time may not provide any benefit due to packaging parasitic inductances and capacitances. 

Due to the high power nature of this application we shall use IGBTs as the example device topology. As 

such, the reduction in switching times may provide benefit due to the “current tail” or delayed turn-off 

characteristic limiting switching speeds rather than packaging parasitics. Modern trench gate IGBTs have 

shown more detriment to breakdown voltage at lower cryogenic temperatures. This reduction in 

breakdown voltage is as high as 70 percent in trench gate IGBTs at approximately 50 K. Although other 

semiconductor devices are available for higher power densities, such as the gate turn-off thyristors 

(GTOs), MOS gated thyristor (MGT), MOS controlled thyristor (MCT), Integrated Gate-Commutated 

Thyristor (IGCT), etc., no research was found that quantified the cryogenic performance of these devices. 

A scaling of device parameters for a selection of IGBTs is shown in Table 6. 

Typical passive components require careful consideration at cryogenic temperatures. Yet, polymer 

film, solid tantalum, and mica capacitors have relatively small changes in capacitance and equivalent 

series resistance and decreased dissipation factors (Refs. 41 and 42). Similarly, high permeability alloy 

cores show little variation over temperature. However, ferrite cores decreased in permeability drastically 

with temperature (Ref. 43).

Recent technical and market progression for hybrid and electrical vehicles has resulted in an overall 

inverter specific power reaching ~17 kW/kg (Ref. 44). A typical mass percentage of equipment is shown 

in Table 7 (Ref. 45). By applying the variation in ratings to the mass percentages of typical lightweight 

inverters for cryogenic operation, one may estimate the mass for generator active rectifiers, generator 

field drives, and motor inverters. Similarly, the component room temperature parameters may be scaled 

for cryogenic temperatures for use in a mechanical and semiconductor hybrid circuit breaker.
NASA/CR—2015-218440 34

TABLE 6.—PUNCH THROUGH (PT), NON-PUNCH THROUGH (NPT), AND TRENCH GATE (TG)

DEVICE PARAMETER VARIATION AT 30 K RELATIVE TO ROOM TEMPERATUR (REFS. 46 AND 47) 

Device PT 

(600 V 250 A)

NPT 

(1700 V 200 A)

TG 

(1700 V 150 A)

Resistance 100% (30 mΩ) 100% (50 mΩ) 100% (50 mΩ)

“Knee” voltage 110% (0.7 V) 200% (0.5 V) 170% (0.7 V)

Turn-off time 50% (400 ns) 15% (2800 ns) 5% (600 ns)

Breakdown voltage 75% (900 V) 60% (2000 V) 35% (1800 V)

Gate capacitance 100% 100% 100%

Switching losses 33% (5.8 mJ) 20% (25 mJ) 20% (33 mJ)

TABLE 7.—AUTOMOTIVE INVERTER COMPONENT

MASS PERCENTAGES (REF. 48) 

Component Mass percentage

Heat exchanger .........................................................................37

Power modules, gate drivers, PWBs.........................................23

Housing.....................................................................................15

Capacitors.................................................................................12

Bus bars ...................................................................................... 7 

Current sensors ........................................................................... 6 

3.6 Cryogenic Power Converter Prototypes

Both high and low power cryogenic power converters are in the early stages of research. One example 

of a low power DC-DC converter was designed, developed, and tested at the University of Manchester

(Ref. 49). The converter was designed as the field controller for a superconducting machine in 2012. The 

machine’s field was superconducting. Basic specifications for the cryogenic DC-DC converter are: 50 V 

output, tested up to 40 A (2 kW), MOSFETs used as switching devices, operated at 77 K, a closed-cycle 

cooler was used, and liquid nitrogen was the cooling medium. 

Higher power converters were prototyped for naval propulsion by MTECH under funding from the 

Missile Defense Agency. The two prototypes, one low voltage at 600 V and one high voltage at 1200 V,

showed increased efficiency, from 97.3 to 99.7 percent, at liquid nitrogen temperature ranges (Ref. 50). 

Yet, the cryogenic efficiency was admittedly difficult to quantify due to the dynamics of the pulse 

waveform and the precision of the test equipment. For the high voltage bridge the increase in efficiency 

was attributed to the greatly reduced turn-off times, thus switching losses.

Due to carrier “freeze out,” or inability of carriers to reach the conduction band below approximately 

40 K for highly doped silicon, large band gap semiconductors must be used if MgB2 is the superconductor 

of choice. Ironically, research into high temperature semiconductor power devices and provides the 

technology for lower cryogenic temperatures (Refs. 51 and 52). Since the highest HVDC conversion 

system is 275 kV, the relative size and cost of the conversion equipment will influence the overall design 

optimization without a hard constraint with respect to voltage (Ref. 53). To derive the weight of the 

conversion equipment, the present strategy is to scale state-of-the-art converters designed for weight and 

cost in the automotive market with respect to each component. The semiconductors will assume IGBTs at 

a maximum of 10X room temperature switching speeds, and a hold off voltage reduction of 30 percent

from room temperature. The passive components will then be a per unit scale of modern mobile inverters 
NASA/CR—2015-218440 35

as a function of state (current for inductors and voltage for capacitors) and frequency. The resulting mass 

will be used for overall system sensitivity analysis. 

3.7 Superconducting Electric Machine Prototypes

While superconducting electric machines are not the focus of this voltage sensitivity study, it is 

beneficial to understand the operating voltages of superconducting machine prototypes. The power 

converters must rectify the generator voltage and invert the DC distribution voltage to control the motors. 

Because of this interface, the power converter voltage sensitivity should also consider electric machine 

voltage operation. 

Several prototypes of partially superconducting electric machines have been developed and tested

(Table 8). Key metrics of these designs are tabulated in Table 9. Additionally, a fully superconducting 

electric machine prototype is being built in 2013 at the University of Manchester. For these prototypes, 

the phase voltages are rated to 2 to 4.5 k. 

3.8 Study Voltage Range Conclusion 

Current installations and prototypes of superconducting cables, SFCLs, electric machines, and 

cryogenic power converters all involve connections to normally conducting and higher temperature 

environments. As a result, the operating voltage of the component is constrained by this connection and 

the surrounding electrical system architecture. These systems range from research prototypes to full-scale 

transmission operation. Based on these installations, the DC distribution voltage range for this TeDP 

superconducting architecture study will be 2.5 to 40 kV with potential extended targets. The lower limit is 

set by the maximum current carrying capacity of existing installations. The upper limit is based on current 

installations of superconducting cables in the power range of the distribution cables for this TeDP 

architecture (≈50 MW). 

An extended range of interest may also be evaluated which would include 250 V at the lower end and 

270 kV on the upper end. These reflect potential future current carrying capacities for DC 

superconducting cables. The upper limit represents the limit for current terrestrial installations. The 

practical final voltage range for the TeDP architecture will be determined by the critical current density of 

the superconductor in the N+3/N+4 time frame in order to generate, convert, and distribute the necessary 

propulsion power. 

System weight will be highly sensitive to the operating voltage selected. Therefore, this deliverable 

was intended to pose a voltage range of interest for in support of future. 
NASA/CR—2015-218440 36

TABLE 8.—CURRENT SFCL INSTALLATIONS AND RESEARCH PROJECTS

Installation 

company 

Date Operating 

voltage, 

kV rms

Rated 

operating 

current,

kA rms

Expected 

max fault 

current, 

kA

Superconductor 

material

Operating 

temperature, 

K 

Rated 

power, 

MW

Cooler Size Type Recovery 

time, 

s 

Ref.

University of 

Manchester, Hyper 

Tech Research

2013 5.5 1.25 2.8 MgB2 20-34 6.875 Gifford 

McMahon

8- by 10-ft skid Resistive 180 54

University of 

Manchester, Hyper 

Tech Research

2012 0.00779 0.283 0.7 MgB2 20-34 0.220457 Scientific 

Magnetics, 

GiffordMcMahon, 

Helium

Resistive 50 55,35

ACCEL/Nexans 2004 12 0.6 BSCCO 2212 bulk 7.2 Resistive 56

Nexans/ASL 2009 12 0.1 BSCCO 2212 bulk 1.2 Resistive 57 

Nexans 2009 12 0.8 BSCCO 2212 bulk 9.6 Resistive 57

Nexans/ASL 2011 12 0.4 BSCCO 2212 bulk 4.8 Resistive 57

Nexans

2011 12 0.560 63 YBCO tape 77 6.72 Nexans 

Open Loop,

LN2

2.5- by 1- by 13-

m, 2.5 tons

Resistive 10 28

Nexans

2013 24 1.005 25.6 YBCO tape 24.12 Gifford 

McMahon, 

LN2

10- by 1- by 3-m Hybrid 28

Siemens/AMSC 2007 7.5 0.3 YBCO tape 2.25 Resistive 58

Siemens/AMSC

2011 138 1.2 63 YBCO tape < 75 K 165.6 Cryomech, 

LN2

8-m long by 

3-m diameter 

40,000 kg per 

phase

Hybrid 

Resistive

15 59

CESI Ricerca 2005 3.2 0.22 BSCCO 2223 tape 0.704 Resistive 60

CESI Ricerca 2006 0.397 0.096 MgB2 tape 0.038112 Resistive 61

ERSE, Sumitomo 2010 9 0.25 30 BSCCO 2223 tape 65 2.25 Stirling BV, 

LN2

3/5- by 2- by 4-

m, 3.8 tons

Resistive 10 62, 

28 

ERSE 2012 9 1 YBCO tape 9 Resistive 62

SuperPower 2004 8.6 0.8 BSCCO 2212 bulk 6.88 Resistive 63

Zenergy 2009 12 1.2 BSCCO 2223 tape 14.4 DC biased 

iron core

64

Zenergy 2011 12 1.2 BSCCO 2223 tape 14.4 DC biased 

iron core

65

CAS 2005 10.5 1.5 BSCCO 2223 tape 15.75 Diode 

bridge

66

Innopower 2007 35 1.5 41 BSCCO 2223 tape 77 90 Open loop, 

LN2

4.2- by 4-m 

diameter, 27 tons

DC biased 

iron core

0.8 67, 

28

Innopower 2010 220 1.36 BSCCO 2223 tape 300 DC biased 

iron core

68

Hyundai 2007 13.2 0.63 YBCO tape 8.316 Resistive 69 

KEPRI/LSIS 2007 22.9 0.63 25 YBCO thin film 71 14.427 Closed 

Loop, LN2

2.5- by 1.2- by 

2.4-m, 1 ton

Hybrid 

resistive

70

Toshiba 2008 6.6 0.072 YBCO tape 0.4752 Resistive 71

TABLE 9.—CURRENT SUPERCONDUCTING ELECTRIC MACHINE PROTOTYPES

Installation 

company 

Date Machine description Rated phase 

voltage, 

kV 

Rated operating 

current,

kA 

Rated 

power, 

MW

Superconductor 

material

Rated 

speed, 

rpm

Cooler Operating 

temperature,

K 

Ref.

University of 

Manchester

2010 Superconducting field winding 

(rotor)

2.71 1.73 7 MgB2 156 20-30 72

AMSC, CAPS, 

ONR

2004 HTS field winding on rotor, 

conventional copper air-core 

winding on stator

2.4 0.715 5 230 73

GE, NREL/DoE 2004 HTS field 4.16 0.255 1.5 BSCCO2223

3600 Closed-cycle GiffordMcMahon, helium74 
NASA/CR—2015-218440 37

4.0 DC Protection Devices

4.1 Introduction

There are several ways to protect superconducting DC electrical systems—dedicated DC protection 

devices, control of rectifiers and inverters, and quench control. Candidates for dedicated DC protection 

devices are the primary focus for this document. An understanding of the characteristics and limitations of 

these devices is required in order to design the electrical system protection and control. However, before 

designing the protection and control, a transient electrical and thermal fault analysis of the complete 

electrical system architecture is necessary in order to determine the protection requirements and decide 

which protection technologies should be used to implement the protection and control design. The 

protection requirements will be defined to meet one or more objectives such as minimizing the protection 

equipment weight or maximizing the efficiency of the protection system. 

The dedicated DC protection devices discussed are electromechanical circuit breakers (EMCBs), 

hybrid circuit breakers (HCBs), and SSCBs. Superconducting fault-current limiters (SFCLs) can also be 

used in conjunction with a circuit breaker in order to further limit the fault current required to be 

conducted and interrupted by the circuit breaker. A detailed discussion of circuit breaker use with SFCL 

is not included in this document but should be studied when determining implementation of the protection 

system. A notable difference between the types of circuit breakers is their operating time as discussed in 

Reference 75 (see Figure 24). For a conventional DC electrical system as well as a superconducting 

system before quenching, the fault energy required to be dissipated by the circuit breaker is reduced when 

the circuit breaker can interrupt the fault current more quickly. Generally, the size and mass of the circuit 

breakers is reduced for lower fault energy dissipation requirements. In the case of a quenched 

superconducting system, the fault current is significantly reduced, so the increased circuit breaker 

operation time may not be as advantageous. However, the energy lost to heat due to the increased 

resistance during quench may have a significant impact on the cooling system size. Additional differences 

discussed for the different circuit breaker technologies include weight and conduction resistance.

Figure 24.—Comparison of Circuit Breaker Operating Time with Time to Peak Current 

for Different Electrical System Architectures (Ref. 75). 
NASA/CR—2015-218440 38

4.2 DC Electromechanical Circuit Breakers

4.2.1 Conventional DC EMCBs

DC mechanical circuit breakers rated for interruption of thousands of amperes have been developed 

for industries such as traction. They employ a mechanical switch and a means to absorb and dissipate the 

fault energy, such as a cold cathode arch chute (Ref. 76). As part of the mechanical circuit breaker, 

applying a magnetic field orthogonal to the arc (by means of permanent magnets, secondary coils, or 

magnetic field due to current) can be used to move the arc from the contact gap into the splitter plates. 

Examples of DC mechanical circuit breakers rated for large DC current interruption are predominately 

developed for traction applications, such as those developed by Hawker Siddeley Switchgear Ltd, 

Sécheron, and GE (Refs. 77, 78, and 79). Figure 25 illustrates the components in an example EMCB.

These DC circuit breakers are developed with different ratings for the rectifier and feeder of a traction 

power substation. Most of these are air circuit breakers with electromagnetic blowout. The electrical 

contact material is silver tungsten carbide. Note that these devices are typically designed to operate from 

–25 to 40 °C and up to 2000 m altitude. These existing designs can be operated at higher ambient 

temperatures up to 55 °C or at higher altitudes, but their operation is then derated. They are traditionally 

not designed to withstand vibrations beyond 0.5 g per 30 s nor for high humidity. Table 10 shows 

characteristics of several EMCBs including weight, conduction resistance, approximate operating time, 

and electrical specifications.

A potential advantage of the mechanical circuit breaker over the hybrid or solid-state circuit breakers 

is lower conduction losses during normal operation, which becomes more significant for higher nominal 

current operation. More information about hybrid circuit breakers is presented in Section 4.3. 

Additionally, the mechanical and hybrid circuit breaker provides a physical separation of conductors 

while the solid-state circuit breaker relies on semiconductor dielectric strength. However, DC mechanical 

circuit breakers take significantly longer to interrupt the fault compared to solid-state energy dissipation. 

Assuming that faults can reliably be detected, discriminated, and the electrical system has not quenched, 

then it is advantageous to interrupt the fault faster in order to reduce the fault energy. 

Figure 25.—Hawker Siddeley Switchgear Ltd Lightning DC Circuit Breaker Mechanical Diagram (Ref. 77). 
NASA/CR—2015-218440 39

TABLE 10.—EMCB DEVICE CHARACTERISTICS 

Device Name Developed by Weight, 

kg

Normal 

conduction 

resistance,

mω

Approx. 

operating 

time, 

ms

Rated 

voltage, 

kV

Normal 

operational 

current 

rating, 

kA

Fault 

current 

rating,

kA

Ref.

Lightning NDC Circuit Breaker Hawker Siddeley Switchgear 580 0.005 100 0.75 4 125 77

Lightning NDC Circuit Breaker Hawker Siddeley Switchgear 600 0.005 100 0.75 6 125 77

Lightning NDC Circuit Breaker Hawker Siddeley Switchgear 600 0.005 100 0.75 8 125 77

Lightning NDC Circuit Breaker Hawker Siddeley Switchgear 620 0.005 100 1.5 4 100 77

Lightning NDC Circuit Breaker Hawker Siddeley Switchgear 640 0.005 20 1.5 6 100 77

Arc Chute 81, HPB45 Secheron 108 Not available 20 0.9 4.5 125 78

Arc Chute 81, HPB60 Secheron 126 Not available 20 0.8 6 125 78

Arc Chute 82, HPB45 Secheron 119 Not available 15 1.8 4.5 80 78

Arc Chute 82, HPB60 Secheron 137 Not available 15 1.8 6 80 78

Gerapid 2607, Arch chute 1x2 GE 120 Not available 20 1 2.6 50 79

Gerapid 2607, Arch chute 1x4 GE 120 Not available 20 2 2.6 35 79

Gerapid 2607, Arch chute 2x2 GE 160 Not available 20 2 2.6 71 79

Gerapid 2607, Arch chute 2x3 GE 160 Not available 20 3 2.6 35 79

Gerapid 2607, Arch chute 2x4 GE 160 Not available 20 3.6 2.6 30 79

Gerapid 4207, Arch chute 1x2 GE 120 Not available 20 1 4.15 50 79

Gerapid 4207, Arch chute 1x4 GE 120 Not available 20 2 4.15 35 79

Gerapid 4207, Arch chute 2x2 GE 160 Not available 20 2 4.15 71 79

Gerapid 4207, Arch chute 2x3 GE 160 Not available 20 3 4.15 35 79

Gerapid 4207, Arch chute 2x4 GE 160 Not available 20 3.6 4.15 30 79

Gerapid 6007, Arch chute 1x2 GE 150 Not available 20 1 6 50 79

Gerapid 6007, Arch chute 1x4 GE 150 Not available 20 2 6 35 79

Gerapid 6007, Arch chute 2x2 GE 165 Not available 20 2 6 56 79

Gerapid 6007, Arch chute 2x3 GE 165 Not available 20 3 6 35 79

Gerapid 6007, Arch chute 2x4 GE 165 Not available 20 3.6 6 Not available 79

Gerapid 8007, Arch chute 1x2 GE 190 Not available 20 1 8 50 79

Gerapid 8007, Arch chute 2x2 GE 210 Not available 20 2 8 50 79

4.2.2 EMCBs Applied to Superconducting Systems

Several instances of superconducting circuit breakers or switches are available in the literature. 

Engineers at the Tokyo Electric Power Company have designed and developed a DC current-limiting 

circuit breaker with a superconducting fault-current limiter (Ref. 80). Their project involved a SFCL with 

a puffer-type DC circuit breaker in liquid nitrogen. They tested the DC circuit breaker design at varying 

voltages and fault currents to determine the mechanical limitations of the circuit breaker with and without 

the puffer operation. Figure 26 and Figure 27 show that the puffer increases the operating voltage, reduces 

the fault interruption time, and increases the fault current that can be interrupted successfully. The fault 

interruption time of approximately 10 to 20 ms is similar to other DC circuit breakers that use air only as 

a dielectric. This study shows that it is feasible to use LN2 as a dielectric for a mechanical circuit breaker. 

The theory for scaling a superconducting switch has been developed by researchers at CERN

(Ref. 81). Researchers there have a need to extract the energy from a superconducting magnet when it 

quenches to limit the heat generated by the event. For their application, the switch connected in series 

with the superconducting magnet can be superconducting or normally conducting. The switch is in 

parallel with a dump resistor. An illustration of an example superconducting switch for this application is 
NASA/CR—2015-218440 40

shown in Figure 28. A superconducting switch may reduce the heat lost due to removal of the interface of 

the superconductor to a higher temperature environment during nominal operation. If the switch operated 

at a higher temperature, the large currents in the circuit would flow from the superconducting to 

conventionally conducting environments. For a superconducting switch, the leads to the higher 

temperature environment only need to conduct during discharge of the superconducting magnet energy. 

The RTAPS architecture has the objective to eliminate conductor leads between superconducting and 

normally conducting environments. Such a superconducting switch could be used as a mechanical circuit 

breaker or a hybrid circuit breaker. As a hybrid circuit breaker, the superconducting switch would be used 

to commutate the fault current to the solid-state circuit which dissipates the fault energy.

Figure 26.—Fault Interruption Time versus Voltage for LN2 DC Circuit Breaker (Ref. 80). 

Figure 27.—Fault Interruption Current versus Voltage for LN2 DC Circuit Breaker (Ref. 80). 
NASA/CR—2015-218440 41

Figure 28.—Basic Circuit for Superconducting Switch to Divert 

Superconducting Magnet Discharge Current to a Dump Resistor. 

The superconducting switch would be a metal matrix or on a metallic substrate. The study in 

Reference 81 discusses switch scaling based on superconducting material, rated nominal voltage and 

current, fault current, and energy dissipation requirements. The minimum mass required for the 

superconducting switch is determined by the energy that must be absorbed by the switch. 

The switch is triggered from a superconducting state to a resistive state by raising the temperature, 

current density, or magnetic field. To maximize efficiency, the temperature is likely to be held as constant 

as possible within the ratings of the cryocooler, and there may not be a means of pulsing a magnetic field 

by the switch. For this scenario, the switch can be designed to transition to a resistive state due to 

increased current density in a similar fashion as a resistive SFCL. The CERN study discusses using 

discharge capacitors to create a large current pulse. 

The study outlines an approximate mass estimate for the switch based on energy, current, voltage, 

superconductor density, engineering current density, resistivity, and energy absorption capability. 

(Ref. 81) This mass estimate is characterized as

Teng

oo s CJ

VIE M

ρ

υσ ≥ 2

2 0

where

Ms mass of switch wire and cable (kg)

Eo energy stored in circuit (J)

I0 maximum current in main circuit (A)

Vo maximum voltage to discharge into resistor (or other energy dissipation device) (V)

υ density of switch cable (kg/m3

) 

σ safety factor to ensure sufficient margin for cable cross section Jeng

I A σ 0 =

Jeng engineering current density limit in switch (A/m2

) 

ρ resistivity of materials in switch conductor (Ω⋅m)

CT specific energy of switch for energy that can safely be absorbed (J/kg)
NASA/CR—2015-218440 42

For an MgB2 superconductor (critical current at 39 K), the study proposes a Cu-Ni conductor for the 

switch matrix. Considering the conductor properties of this switch, one can use the following estimates 

for the parameters (Ref. 81): 

υ = 8×103 kg/m3 

σ = 1.5 

Jeng = 109 A/m2 (for 2035 time frame)

ρ = 50×10–8 Ω⋅m (stainless steel and copper-nickel matrix)

CT = 40×103

 J/kg (for operation up to 250 K) 

Given these assumptions, the mass of the superconducting switch can be estimated based on the 

energy dissipation requirement, current, and voltage. To size a superconducting switch on one of the four 

main TeDP distribution lines connecting each generator to a feeder bus (rated for 50 percent of minimum 

takeoff power = 12.5 MW), and assuming an interruption time of 1 ms (for a hybrid circuit breaker 

response time), the energy that must be dissipated by the switch is 12.5 kJ. This is merely an 

approximation of the energy for this example. A transient analysis of the fault current and inductance for 

faults at various locations on the architecture is required to determine the energy dissipation requirement 

and the appropriate protection system fault detection and response time in order to minimize the mass of 

the protection equipment or meet some other objective. Given this assumption, the mass of the switch can 

be estimated as:

00 4

29 8 3

23 0 3

2

2 0

10004.1

10401050)10(

102.11 5.1108

M VI

VI M

CJ

VIE M

s

o s

Teng

oo s

−

−

×≥

××××

×××××

≥

ρ

υσ ≥

For the fixed power of 12.5 MW, the switch mass is estimated as 0.34 kg. However, the maximum current 

and voltage will differ from the steady-state power requirement. Figure 29 shows the switch mass 

estimate for a voltage range of 2.5 to 40 kV and current range of 280 to 4480 A. The switch mass ranges 

from 0.084 to 1.344 kg. 

If the fault interruption time is longer as is typical for electromechanical circuit breakers (10 ms), then 

the fault energy increases proportionally (125 kJ). For the fixed power of 12.5 MW, the switch mass for 

this increase in fault energy increases more than three times to 1.06 kg. 

4.3 Hybrid Circuit Breakers

There are many different topologies of HCBs. In essence, each includes a mechanical switch to 

commutate the current into the solid-state device that is in parallel with the switch. The solid-state device 

is used to dissipate or store the energy in the arc. The mechanical switch must be rated to carry the 

nominal and fault current and should be designed to have a low on-state resistance. The solid-state 

devices are used for current conduction for a short time and can be rated to just the interrupted fault 

current. Some topologies are depicted in Figure 30, Figure 31, and Figure 32. 
NASA/CR—2015-218440 43

Figure 29.—Superconducting Switch Mass Estimate for Fixed Energy Dissipation Requirement. 

Figure 30.—Simplified Schematic of HCB with IGCTs Prototype (Ref. 82). 
NASA/CR—2015-218440 44

Figure 31.—Simplified Schematic of HCB using IGCTs and Metal-Oxide Varistor (Ref. 83). 

Figure 32.—Schematic of HCB for HVDC using IGBTs (Ref. 84). 

The circuit breaker in Reference 82 consists of several conducting paths in parallel and a bypass 

switch (BPS). The BPS was rated for 70 kA nominal current and 17.5 kV nominal voltage and is opened 

by a pneumatic actuator. The BPS arc voltage is low for the paper’s application in a superconducting 

magnet system. This posed problems for the current commutation from the bypass switch to the circuit 

breaker. Three paralleled IGCTs with snubbers and voltage clamps were used to form the circuit breaker 

due to their controllability at turn-on and turn-off and ability to interrupt large currents. Each conduction 

path of this circuit breaker is composed of an inductor (required for IGCT turn-on protection to limit the 

rate of current rise), diode, IGCT, and current measurement device. Both studies in References 83 and 84

investigate and use Thomson drives for fast opening times of the mechanical switch. 

While mass data is not published for these HCBs, the existing IGCT and IGBT component mass as 

well as other characteristics, such as conduction resistance, are available. Approximations of the mass for 

the other components in the hybrid circuit breaker can be made in order to estimate the hybrid circuit 

breaker weight. Available data for several hybrid circuit breaker implementations and IGBT and IGCT 

components is tabulated in Table 11. Additionally, the table includes a solid-state circuit breaker 

prototype for comparison with the HCBs. This data is developed for IGBT or IGCT operation at 25 to 

125 °C. An understanding of the semiconductor at cryogenic temperatures (4 to 70 K) is required in order 

to estimate conduction resistance, blocking voltage, and turn-on and turn-off times for cryogenic 

electrical system protection. This discussion is included in Section 3.5. 
NASA/CR—2015-218440 45

TABLE 11.—HCB, SSCB, AND IGBT AND IGCT COMPONENT DATA

Device name Developed by Use Weight, 

kg

Conduction 

resistance 

at rated 

current, 

mΩ

Peak offstate 

current, 

mA 

On-state 

voltage, 

V 

CB turn-off 

(component 

turn-on) 

operating time,

ms 

Rated 

blocking 

voltage 

rating, 

kV

Normal 

operational 

current 

rating, 

kA

Fault 

current 

rating, 

kA

Ref.

Hybrid DC CB European Atomic 

Energy 

Community 

(EUATOM) 

Quench 

protection of 

superconducting 

magnets

Not 

available

75.00 Not 

available

24 8 2.8 4 10 82

DC hybrid CB with 

ultra-fast contact 

opening and IGCTs

ABB Railway electrical 

system

Not 

available

1.00 Not 

available

5 0.3 2.5 4 5.7 83

Hybrid HVDC CB ABB HVDC VSC 

Protection

Not 

available

Not 

available

Not 

available

Not 

available

0.2 80 2 8.5 84

Solid-State CB with 

Six Series-Connected 

IGBTs

Diversified 

Technologies

Naval power 

system

27.2 11.25 10 9 0.0045 10 0.8 1 85,

86

ABB HiPak IGBT 

Module 5SNA 

1200G450300

ABB Hybrid DC circuit 

breakers 

1.76 2.17 0.0005 2.6 0.00098 4.5 1.2 2.4 87

IGCT module - 5SHX 

26L4520 (Reverse 

Conducting IGCT)

ABB Hybrid DC circuit 

breakers 

2.9 1.27 50 2.6 0.007 2.8 1.01 2.2 88

From Table 11, there are several differences among the technologies to note. The conduction 

resistance for IGBTs is higher than for IGCTs. This may become significant at higher fault current levels 

but is dependent on the component operation at cryogenic temperatures. The IGBT component weight is 

less than the IGCT weight, where the IGBT has comparable fault current interruption capability but 

higher blocking voltage rating. This may be beneficial to reduce the solid-state component weight when 

using several components in series or parallel. The IGBT turn-on time is faster than the IGCT time, but at 

cryogenic temperatures the times may differ. 

4.4 Solid-State Circuit Breakers

There are several solid-state circuit breaker designs, but few with published mass and efficiency data 

and testing data for higher voltage and current operation. One example with mass estimate and transient 

opening and closing data was developed by Diversified Technologies for naval power systems (Ref. 85).

They developed and tested a solid-state circuit breaking using six 4.5 kV IGBTs (CM900HB-66H) 

connected in series. Figure 33 and Figure 34 illustrate the physical layout and the simplified SSCB 

diagram. The published mass estimate for the solid-state circuit breaker was 27.2 kg (60 lb). Looking up 

the Mitsubishi Electric high-voltage IGBT data sheets, each IGBT weighs 1.5 kg. For six IGBTs, the 

IGBTs account for approximately 9 kg (33 percent of total weight). That leaves an additional 18.2 kg 

(67 percent) for the additional structural components as well as snubbers and other circuit components. 

Additional discussion about cryogenic operation of IGBTs and characteristics of IGBTs is included in 

Section 3.5. 
NASA/CR—2015-218440 46

Figure 33.—A 10 kV, 800 A IGBT Solid-State Circuit Breaker Mechanical Layout (Ref. 85). 

Figure 34.—Simplified Diagram of Example Solid-State Circuit Breaker (Ref. 85). 

Referring to Table 11, there are several notable differences between HCBs and SSCBs. The 

conduction resistance of SSCBs is generally higher than that for HCBs due to the lower resistance of the 

mechanical switch for HCBs. As mentioned previously, the turn-on (current interruption) time for the 

SSCBs is significantly faster than for HCBs in large part due to the operation of the mechanical switch for 

the HCBs. Practical implementation of the protection and control system that enables fault detection and 

discrimination fast enough to utilize the fast interruption time of SSCBs is difficult. The transient fault 

response for the electrical system architecture will determine whether or not the protection system weight 

can be significantly reduced by using SSCBs and their associated faster fault interruption time. 
NASA/CR—2015-218440 47

TABLE 12.—DC PROTECTION DEVICE SUMMARY

Characteristic/Device EMCB HCB SSCB

Operating time – o + 

Weight – o + 

Conduction resistance + o – 

Cryogenic operation o + + 

4.5 Summary

The major properties of DC protection devices including weight, conduction resistance, fault current 

interruption capability, and fault interruption time were presented based on available data. Further 

assessment can be done to estimate EMCB, HCB, and SSCB weight for varying fault current interruption 

and blocking voltages based on IGBT or IGCT component weights and estimating additional CB 

component weights such as inductors, contactors, and arresters. The summary of the discussion in this 

section is shown in Table 12. The + symbol indicates a benefit, o indicates a neutral or minor benefit, and 

– indicates a negative attribute for application to a superconducting TeDP protection system. 

For the sensitivity and dynamic modeling, SSCB were used to eliminate fault conditions quickly and 

had a compact package. If the devices operate early or late in the fault condition the amount of fault 

current that it is required to interrupt is decreased significantly which will reduce the size of the circuit 

breaker. The longer a fault persists on the network increases the amount of energy that the cryocooler will 

have to absorb increasing the size and mass of the system. These two reasons imply that interrupting a 

fault early in the cycle is the preferred option to reduce size and weight. The coordination between the 

superconducting fault current limiter and circuit breakers can be explored using the dynamic model. If the 

superconducting fault current limiter can restrict the peak of the fault current to non-damaging levels than 

slower circuit breakers may be used but the prolonged fault condition will result in a larger cooling 

system.
NASA/CR—2015-218440 48

5.0 Component Sensitivities and Sensitivity Modeling

5.1 Component Sensitivity Overview

The impact of operating voltage at the system level is dependent on its integrated effects on all system 

components. In order to determine the sensitivity of the TeDP architecture to operating voltage, mass and 

efficiency sensitivities of each system component must be characterized. Parametric sensitivity models 

were developed for TeDP components. 

This section outlines the methods employed for determining component voltage sensitivity. Important 

assumptions are highlighted and the driving factors behind sensitivity trends are discussed. Component 

sensitivity models reviewed in this section are:

• Converters 

○ Unidirectional current source converter 

○ Bidirectional current source inverter

• Superconducting cables 

• Superconducting magnetic energy storage 

• Solid-state circuit breaker

• Superconducting fault current limiter 

The component I/O was determined considering the overall system requirements, system sensitivity 

variable, component design parameters and interactions between components and systems. The 

relationships are indicated via the colored arrows in Figure 35. The weight and efficiency for each 

component is determined as a function of the input variables and internal assumptions. Additionally, the 

component losses and temperature at which the heat is removed determines the heat quality factor 

illustrated in Figure 35. 

Component specific parameter diagram will be given with the discussion of each TeDP component 

model to illustrate the parameters used in its model. 

The power requirement and component count for each architecture discussed in Section 2.0 is provided in

Table 13. This component breakdown is modified from the breakdowns presented in the August 2012 

Rolls-Royce RTAPS final report (Ref. 89). The architectures considered here have 16 propulsors instead 

of 14. The nominal power requirements for the components are scaled accordingly. The nominal current 

rating for the each grid component is determined by the power rating and the operating voltage for each 

component. 

The max operating current for each device requires more information regarding the manner in which 

the protection equipment is configured to mitigate overcurrents during electrical system failures. The 

identification of the sizing current requirements will be discussed in the subsequent sections. 

Table 13 gives a high level decomposition of the architecture. However, component sizing requires a 

decomposition of the architecture to lower levels of abstraction. Each component is an assembly of 

subcomponents which contribute to the overall weight and component efficiency. 
NASA/CR—2015

-218440 49

Figure 35.—Overall System Parameter Diagram with Legend for Arrow Colors.

Architecture Systems

Generator/Motor

Cable

Rectifier/Inverter

Energy Storage

Protection Systems

Circuit 

Breaker

SFCL

Bus Tie

System Parameter Diagrams

Component 

Weight 

[Arch]

Component 

Efficiency 

[Arch]

Heat Quality 

[Arch]

Quench 

Current Margin 

[Arch]

Temperature 

[Arch]

Superconductor 

Type [Arch]

Field Strength 

[Arch]

System 

Requirements 

Management

Voltage [Arch]

Nominal Rated 

Current [Arch]

Topologies [Arch]

Rated Power [Arch] Fault Response Time Constant [Arch]

Max Operating

 Current [Arch]

R, L, C

Protection 

Component 

Weight 

[ProtSys]

Protection 

Component 

Efficiency 

[ProtSys]

Nominal Rated 

Current [Arch]

Voltage [Arch]

Voltage [Arch]

Fault Current 

Ratio [Arch]

Max Operating

 Current [Arch]

Legend

= Fixed according to architecture= System-wide input variable by user= Component control= Output= Input or output within systems (architecture or protection)
NASA/CR—2015-218440 50

TABLE 13.—ARCHITECTURE COMPONENT BREAKDOWN

Components Arch1:

Baseline

Arch2:

Inner bus tie

Arch3:

3-bus multi-feeder

Arch4:

Cross-redundant 

multifeeder

Arch5:

4-bus inner bus tie, 

multifeeder

Count Rating, 

MW 

Count Rating, 

MW 

Count Rating, 

MW 

Count Rating, 

MW 

Count Rating, 

MW 

Electric 

Machines

Generator 

4 11.19 4 14.91 4 11.19 4 11.19 4 11.19

 2 7.46 

Motor 16 2.80 16 1.86 16 1.60 16 1.60 16 1.60

Converter

Unidirectional 

AC/DC Converter

4 11.19 2 14.91 4 11.19 4 11.19 4 11.19

 2 7.46 

Bidirectional DC/AC 

Inverter 16 2.80 16 1.86 16 1.60 16 1.60 16 1.60

DC/DC Converter 

for SMES

4 11.19 2 14.91 4 11.19 4 11.19 4 11.19

 2 7.46 

Cables

Transmission

4 

11.19

(2- by 30-m, 

2- by 40-m) 

2 

14.91

(1- by 30-m, 

1- by 40-m) 

4 

11.19

(2- by 30-m, 

2- by 40-m) 

4 

7.46 

(2- by 30-m, 

2- by 40-m) 

4 

11.19

(2- by 30-m, 

2- by 40-m) 

 2 

7.46 

(1- by 30 m, 

1- by 40 m)

 4 

3.73 

(2- by 30-m, 

2- by 40-m) 

 

Feeder

16 2.80

(16- by 5-m) 16 1.86(16- by 5-m) 16 1.60(16- by 5-m) 16 1.60(16- by 5-m) 16 1.60(16- by 5-m) 

 16

1.20

(16- by 5-m) 16 0.80

(16- by 5-m) 16

1.20

(16- by 5-m) 

Breakers

AC

4 11.19 4 14.91 4 11.19 4 11.19 4 11.19

 2 7.46 

16 2.80 16 1.86 16 1.60 16 1.60 16 1.60

DC

8 11.19 4 14.91 8 11.19 8 7.46 8 11.19

 4 7.46 8 3.73 

32 2.80 32 1.86 32 1.60 32 1.60 32 1.60

 32 1.20 32 0.80 32 1.20

 1 7.46 1 9.60

SFCL

AC 

4 11.19 2 14.91 4 11.19 4 11.19 4 11.19

 2 7.46 

16 2.80 16 1.86 16 1.60 16 1.60 16 1.60

DC 

4 11.19 2 14.91 4 11.19 4 7.46 4 11.19

 2 7.46 4 3.73 

 1 7.46 1 9.60

Energy 

Storage SMES

4 11.19 2 14.91 4 11.19 4 11.19 4 11.19

 2 7.46 
NASA/CR—2015-218440 51

Figure 36.—Circuit Diagram for One Generation, Rectification, Distribution, Inversion, to Propulsor Motor Load Line. 

In Addition, an SMES Energy Storage Device and Connections to Other Feeders and the Bus Tie Connection to a 

Secondary Distribution Line are Illustrated.

Figure 36 illustrates this decomposition to a lower level of abstraction for the section of the 

architecture indicated in the black box. 

It should be noted that the decompositions illustrated in this figure assume specific converter 

topologies. This figure illustrates a unidirectional voltage source converter for the AC/DC converter as 

well as a bidirectional voltage source converter for the propulsor drive. Additionally, the DC/DC 

converter for the SMES is simply representative. Discussion on alternative converter configurations and 

the specific sensitivity models generated for this study can be found in Section 5.2. 

5.2 Rectifier and Inverter

This section describes the models of all of the rectifiers and inverter/rectifiers used in this 

architecture. First, the major inputs, outputs, and internal parameters for a system study using one or more 

of these power converters are described. Then, the selected topology and other considered topologies are 

discussed. The rectifier model and its major components are described in detail with significant model 

trends presented and discussed. The major components used in the converter models are also used in the 

solid-state circuit breaker model. Lastly, the bidirectional model trends are presented and discussed. 
NASA/CR—2015-218440 52

5.2.1 Model Overview

5.2.1.1 Parameter Diagram 

The system-level power converter model is described by the parameter diagram shown in Figure 37. 

For the fixed architecture, the rated power is fixed and is indicated as an input to the model. Inputs to the 

model that can be varied are the DC voltage, electrical AC frequency, and power factor. The controls 

indicated in green are the converter topology, switch type, switching frequency, DC ripple voltage, and 

DC ripple current. For this study, the topology and switch type are fixed, but the model could be 

expanded to include system studies of different topologies. Outputs of the model indicated in blue are the 

converter mass, heat quality (losses), efficiency, and optionally harmonic content. At this stage, the 

harmonic content will not be computed. The intermediate outputs that are sent to the protection system to 

determine the protectiond device requirements are the converter fault current rating, current interruption 

time, equivalent capacitance, and equivalemtn arm inductance. Additional intermediate outputs from the 

converter to other subsystems such as the cable model are the AC voltage and AC current as determined 

by the converter specified power, DC voltage, and power factor. These identified parameters were used to 

structure the model and determine interactions with other subsystems. 

5.2.1.2 Converter Topology 

Several converter topologies were brainstormed before choosing a specific topology for which to 

model. With the converter AC conductor interface described by a three-phase system and the DC 

conductor interface described by a bipolar DC system with ground point, neutral (or mid) point clamped 

topologies were considered. This DC bus configuration is also called a three-level (or generically 

multilevel) configuration since it involves defined and controlled positive, grounded, and negative 

potentials. The traditional voltage-source mid-point clamped topology as shown in the system circuit 

diagram of Figure 37 was considered. This voltage-source converter (VSC) topology is widely studied 

and is the most common implementation of low and high power inverters and rectifiers. However, while 

studying and considering the design of this superconducting DC microgrid protection system (Ref. 90), 

the current-source converter (CSC) topology was also considered (Ref. 91). The CSC topology and 

switch module options will be discussed further in Sections 5.2.2 and 5.2.4. Figure 38 shows several 

switch module circuit options for use with the VSC. Each of these circuits would be used where each 

yellow rectangle is drawn in the VSC of Figure 36. Figure 38(a) shows a GTO used as the switching 

device with an antiparallel diode (Ref. 92). The GTO can be turned on and off (unlike an SCR) and has 

long turn off times (on the order of 15 μs). Figure 38(b) shows an IGBT used as a switching device with 

an Emitter Turn-Off Thyristor (ETO) and Metal Oxide Varistor (MOV) in parallel (Ref. 93). The ETO 

provides the switching module the ability to limit or interrupt current, and the MOV is used for 

overvoltage protection of the IGBT and ETO. Figure 38(c) shows a simple IGBT with antiparallel 

freewheeling diode. This switch topology cannot limit current and lacks overvoltage protection. 

The VSC switch chosen impacts the converter current-limiting capability. The VSC switches are 

controlled as a converter to regulate the specified DC bus voltage. During the onset of a DC bus pole-topole fault, the large DC bus capacitor current is discharged into the short circuit (Ref. 94). If the fault 

current magnitude can be limited, then the fault current withstand rating of the affected protection zone 

can be reduced, which potentially reduces the mass of the components. VSCs with current-limiting switch 

modules such as that of Figure 38(b) or CSCs. The CSC topology is essentially the dual of the VSC 

topology. Where capacitors are used to store energy for a CSC and regulate voltage, inductors are instead 

used for a CSC to store energy and regulate current. This control inherently limits current. 
NASA/CR—2015-218440 53

Figure 37.—Parameter Diagram for Power Converter. 

Figure 38.—Potential Switch Topologies for a Voltage Source Converter. 

Not Modeled in Current Study; Current-Source Converters Were Selected 

To Be Modeled.

Additionally, if the duration from the onset of the fault to the peak fault current magnitude be 

lengthened, this allows more time to detect the fault, discriminate the fault location, and interrupt and 

isolate the faulted section of the network. This can be accomplished with current control or the addition of 

inductance or resistance in the circuit such as with the use of a fault-current limiter. 

Converter topologies that can be controlled to limit and interrupt current may reduce the mass of the 

protection devices. However, current-limiting converters make the detection and discrimination of a short 

circuit fault more difficult. Algorithms and analysis of the detection and discrimination of these faults for 

compact DC networks are discussed in Reference 95. 

A comparison of VSCs and CSCs is listed in Table 14. For this system model, the CSC topology was 

chosen over the VSC due to the inherent current-driven nature of a superconducting network enabled by 

high current density of superconducting material. Additionally, it is desired to study the impact of the 

current-limiting capability of the CSC on the protection device weight and system efficiency. 
NASA/CR—2015-218440 54

TABLE 14.—COMPARISON OF VSC AND CSC (REFS. 96, 97, AND 98) 

Comparison Current-source converter Voltage-source converter (dual of CSC)

Control Current, voltage varies with power Voltage, current varies with power

Reverse power flow Voltage changes polarity Current changes polarity/direction 

Short-circuit fault tolerance Tolerant Susceptible, requires short circuit interruption 

protection

Open-circuit fault tolerance Susceptible, requires emergency current path Tolerant 

Efficiency Less efficient More efficient 

Harmonics More harmonics, so larger filter required Fewer harmonics 

Switch bidirectional rating Bidirectional voltage blocking Bidirectional current blocking 

Semiconductor rating Semiconductors rated for full AC voltage Semiconductors rated for full AC current 

As part of the system study, additional converter models could be developed (VSC and CSC), and the 

sensitivity of the converter topology and impact on system weight and efficiency could be assessed. 

5.2.2 Source-Side Converters

With the assumption that the current-source converter is the appropriate converter topology for the 

superconducting power distribution system, there are several configurations of the CSC depending on the 

direction of power flow. The higher power CSCs that convert the generator AC voltage to distribution DC 

voltage only require power flow in one direction (i.e., AC to DC). This simplifies and reduces the 

component count of the switch modules so that the switch modules are designed to only carry current in 

one direction. For a CSC, this is a half bridge configuration of the switching devices. The lower power 

CSCs that normally convert the distribution DC voltage to motor AC voltage may require power to flow 

in both directions (i.e., DC to AC or AC to DC). Such a scenario where power flows from the AC to DC 

side of the converter is in the case of the propulsor “windmilling” and turning the electric machine so that 

it acts like a generator and can send power to the SMES or other loads. For this scenario, the switch 

modules require more components so that the converter can carry current in either direction. For a CSC, 

this is a full bridge configuration of the switching devices and requires twice the number of switching 

devices compared to a half bridge. 

First, the higher power unidirectional current source rectifier, scaling equations, and model trends will 

be presented, followed by the same discussion of the bidirectional current source converter. The converter 

current and voltage rating equations apply to both the unidirectional and bidirectional converters, but the 

component voltage and current ratings for the topologies slightly differ. These differences are indicated 

by referring to the switch modules as half bridge (unidirectional) or full bridge (bidirectional) 

configurations. The scaling of each of the converter individual components (IGBT and diode, capacitor, 

inductor, and housing) are the same for both topologies. 

5.2.2.1 Unidirectional Current Source Rectification

The topology of the unidirectional current source rectifier is shown in Figure 39. The major 

components of this converter are the switching modules indicated by the yellow rectangle, capacitors, and 

clamping diodes. For nomenclature, the series or parallel configuration of the switching modules 

connected from a phase to the positive or negative terminal is called an arm. 

Within a switching module, several unidirectional topologies can be chosen. Some options considered 

are shown in Figure 40. Note that major differences between these current-source and voltage-source 

converter switch modules (shown in Figure 36) are that the devices are aligned in series, inductors are 

required to maintain a current path and store energy, and the IGBTs are paired with series diodes rather than 
NASA/CR—2015-218440 55

antiparallel freewheeling diodes. Both CSC and VSC switch modules may require a device for overvoltage 

protection such as an ETO or other thyristor (Figure 33(b)) or press pack diodes (Figure 40(c)). Each switch 

module may also require a capacitor rather than a lumped arm capacitor as shown in Figure 39. The trends 

developed from these models are derived from a lumped arm capacitor. Further studies could be done to 

assess the capacitor mass for a capacitor per each switch module rather than a lumped capacitor. The 

topology boxed in red (Figure 40(c)) was chosen for this study due to the suggestion of the use of press pack 

diodes (Ref. 99) and the ease of scaling and modeling them over thyristors. 

Figure 39.—Unidirectional, Current-Source Converter Topology 

Selected for the Power Generator Rectifier. The Yellow Blocks 

in This Diagram Represent Potential Variations in the Switch 

Topologies. Note Also That the Arrow Indicates the Direction of 

Current Flow.
NASA/CR—2015-218440 56

Figure 40.—Half Bridge, Unidirectional, Current-Source Converter Semiconductor Switches. Each 

of These Blocks Can Be Connected in Parallel to Subsequent Blocks. Topology C (Press Pack 

Diodes for Overvoltage Protection) Was Selected for the Switch Topology for the AC to DC 

Converter over Topology B (Thyristor for Overvoltage Protection) (Refs. 100 and 101). 

5.2.2.2 Governing Equations

5.2.2.2.1 Converter Sizing Model

The overall converter mass and loss estimate model is used to derive the subcomponent ratings. 

Based on these ratings, the subcomponent weight and losses are estimated. These subcomponent weights 

and efficiencies are then used to estimate the overall converter mass and efficiency depending on the 

number of subcomponents and their configuration. The subcomponents for the power electronic devices 

(unidirectional rectifier, bidirectional rectifier/inverter, solid-state circuit breaker, and bidirectional DCDC converter) are an inductor, capacitor, and IGBT with diode (series or freewheeling). The solid-state 

circuit breaker also includes a varistor whose weight is estimated proportionally to the IGBT weight. 

Further detail to estimate the mass and normal operation efficiency of a varistor or other overvoltage 

protection device could be added to the model to build a cryogenic scalable varistor dependent on voltage, 

current, interruption times, and cryogenic operation of the solid-state circuit breaker. 

The overall converter mass and loss estimate is driven by the specified rated DC power, nominal DC 

rated voltage (pole to pole), AC side frequency, and switching frequency. Given these specifications, the 

converter model calculates several additional ratings. The rated DC current is simply calculated as:

ptpdc

dc

dc V

P I ,

=

Where Pdc the specified is rated DC power and Vdc,ptp is the nominal DC rated voltage (pole-to-pole). 

The rectifier/inverter nominal peak AC current for a given rated DC current is described by the 

equation 

= cosθ

2

1 mIIac dc (Ref. 102) 

Where m is the converter modulation index (between 0 and 1) and cos θ is the power factor (assuming 1 

for calculations throughout this report). 
NASA/CR—2015-218440 57

The rectifier/inverter nominal peak AC voltage for a given rated DC voltage is described by the 

equation: 

ac Vdc

m

V

θ = cos

1

3

22 (Ref. 103) 

These equations are ideal assuming that the input power equals the output power. The AC current or 

voltage ratings could be modified to include the converter losses through increasing the voltage or the 

current. The three-phase real power is calculated by the standard equation: 

θ = cosθ

2

3

3 acacIVP

Other inputs that can be varied that also effect the overall converter mass and lost estimate are the 

operating temperature, DC ripple voltage magnitude, and DC ripple current magnitude. The diode losses 

are calculated as a function of operating temperature. The model could also be expanded to calculate 

IGBT losses, inductor mass and losses, capacitor mass and losses, and heat exchanger as a function of 

operating temperature, but those dependencies are not currently included in the model. The DC ripple 

voltage magnitude is used to determine the capacitance requirements to support the DC bus voltage, and 

the DC ripple current magnitude is used to determine the inductance requirements for each switching 

module. 

The capacitance requirement for the specific current-source converter topology is not analytically 

known but can be approximated using standard topology requirements. For a standard three-phase AC to 

two-level DC full-wave diode rectifier, the rectifier capacitance for voltage regulation is approximated by:

dcac

dc

eq Vf

I C

∆ = 2 (1) 

Where fac is the AC frequency and ∆Vdc is the peak-to-peak DC voltage ripple magnitude. Note that this 

topology differs in that it is actively controlled with IGBTs and the DC side has three voltage levels (midpoint clamped), so this capacitance definition is approximate across the pole-to-pole DC bus. It is 

apparent from this equation that the AC frequency influences the capacitance; a higher AC frequency 

lowers the capacitance required for the same DC bus voltage ripple and DC bus current. 

Similarly, the inductance requirement for this CSC topology is not analytically known but can be 

approximated from standard topology theory. For the same full-wave diode rectifier with inductive 

filtering on the DC side, the DC ripple current is approximated by the following phasor equation at the 

ripple frequency ω = 2π(2fac):

( RLCLjR )

V

I ac DC 2 ,ripple 3

~ 22

ω−ω+π = (Ref. 104) (2) 

Where R is the DC pole-to-pole resistance, L is DC inductive filter, and C is the equivalent DC capacitive 

filter. This equation is adapted to the find the equivalent inductance for a specified DC current ripple by 

assuming that each arm for a three-phase AC side contributes to approximately one third of the total DC 

current ripple. With this assumption and using the quadratic formula, the arm equivalent inductance is 

computed. 
NASA/CR—2015-218440 58

The subcomponent ratings are determined by the overall converter specified voltages and currents. 

The IGBT and diode, inductor, and capacitor subcomponent masses and losses are estimated for these 

ratings and then totaled according to the converter topology to estimate the overall converter loss and 

mass. The following sections describe each subcomponent and how it is scaled.

5.2.2.2.2 IGBT and Diode Sizing Model

The cryogenic operation of the IGBTs and diodes is scaled from room temperature IGBT and diode 

data based on cryogenic testing and research of these devices (Refs. 105, 106, and 107). For more 

information on the cryogenic semiconductor device research summarized for this study, see Section 3.5. 

The cryogenic scaling of the IGBT devices is derived from cryogenic IGBT test results. Proportional 

scaling is used based on these results discussed in Section 3.5 and Table 6. These results are based on 

tests 50 to 300 K. The IGBT cryogenic scaling factors used in each of the models using this device is 

summarized in Table 15. These scaling factors can be modified and thus improve the model estimates as 

further research and test results are completed to better approximate the IGBT performance as a function 

of temperature, voltage, current, and other factors. 

The results from the dissertation research (Ref. 108) were used to approximate the cryogenic behavior 

of diodes as a function of temperature. The turn-on voltage increases approximately linearly as 

temperature falls at approximately the rate of 1.6 mV/K for a 1700 V Dynex diode module tested 50 to 

300 K in 25 K increments. Based on this result, the on-state voltage is estimated by

VV ( ) T K

onD

cryo

onD += 3000016.0 − 300

,,

which is approximate for a temperature range of 50 to 300 K. 

The diode on-state resistance from Reference 109 indicates an approximate linear increase from 200

to 300 K, and a linear but lower slop from 100 to 200 K. Below 100 K, the results show significant 

increase in on-state resistance likely due to carrier freeze out. An approximate linear fit from this data 

yields a slope of 0.01212 mΩ/K for 100 to 300 K. Using this slope, the diode on-state resistance can be 

approximated as a function of temperature by the equation: 

10212.1 ( ) 300 300 5 ,, −⋅+= − RR T K

onD

cryo

onD

TABLE 15.—IGBT CHARACTERISTIC SCALING FOR CRYOGENIC TEMPERATURES

IGBT characteristic

Scaling factor 

cryo/room 

temperature

Comments

Nominal current, A 1 

Over current, A 1 

Nominal blocking voltage, kV 0.5 35 to 75% reduction at lower temperatures

Conduction voltage drop, V 1 

Time to turn off, μs 0.3 5 to 50% reduction at lower temperatures 

Time to turn on, μs 1 

Turn off energy, J 0.25 20 to 33% reduction at lower temperatures 

Turn on energy, J 0.25 20 to 33% reduction at lower temperatures 
NASA/CR—2015-218440 59

For cryogenic diode operation, the tests from Reference 110 indicated an approximate 20 percent 

decrease of breakdown voltage at 50 K compared to 300 K. IGBTs indicated a 35 to 75 percent reduction 

in breakdown voltage for temperatures down to 30 K. It is assumed that the reduction in breakdown 

voltage is approximately the same for IGBTs and diodes. For IGBTs and diodes, the breakdown voltage 

is used in part to determine the series configuration for the rated converter voltage. For the diodes, the 

series configuration is the only use of the breakdown voltage. 

To estimate the cryogenic scaling of the reverse recovery energy for the diode, the results for the 

reverse recovery time as a function of temperature are used (Ref. 111). The reverse recovery energy is

related to the reverse recovery time. This study makes the assumption that the recovery energy is directly 

proportional to the relationship of reverse recovery time and temperature. For 100 to 200 K, the reverse 

recovery time was shown as constant at 300 ns (61 percent of 300 K time), and then increased linearly 

from 200 to 300 K with a slope of 1.63 ns/K (0.3 percent/K). These percentages are used to scale the 

300 K diode reverse recovery energy as a function of temperature by the equation:

( ) 





>−+

≤≤ = E KTTE KTK

E

K rr

K

cryo rr rr 200,300003.0

100,61.0 200

300

300

The overall converter model specifies the cryogenic device nominal current and blocking voltage 

ratings. From these specifications, the IGBT mass, power loss, and current interruption time are estimated 

for the device at the specified device ratings. The existing IGBT ratings for ABB, Infineon, and 

Mitsubishi devices were tabulated, and trends from these devices were developed in order to develop a 

scalable IGBT within the current and voltage ratings of the known devices. Trends were developed for 

IGBTs with nominal current in the range of 200 to 1500 A and nominal blocking voltage in the range of 

1.2 to 6.5 kV. This tabulated data is included in Appendix B as well as in all of the models using IGBTs. 

Because these trends are within a current range of 200 to 1500 A and voltage range of 1.2 to 6.5 kV, 

modules of IGBTs (half bridge for the unidirectional converter and full bridge for the bidirectional 

converter) are configured in series and/or in parallel to size the IGBTs within the current and voltage 

range for which data is available. If the current rating of the overall converter is higher than the maximum 

current for IGBT data, then two or more IGBT modules are connected in parallel to divide the overall 

converter current into the multiple modules and thus avoid extrapolating the IGBT sizing trends. 

Analogously, if the voltage rating of the overall converter is higher than the maximum voltage for IGBT 

data, then two or more IGBT modules are connected in series to divide the overall converter voltage. On 

the minimum rating of the IGBTs, the minimum voltage is within the minimum converter rated voltage 

scaling, but the minimum IGBT current data is greater than the minimum converter rated current for the 

lower powered bidirectional converters at higher voltages. The trends presented in this report thus are 

extrapolating the IGBT scaling for these conditions. An improvement to the models to avoid extrapolation 

of the IGBT trends is to gather IGBT data at lower current ratings but similar voltages and add this data 

and corresponding trends to the IGBT scaling model. 

The rated blocking voltage across the IGBT module is:

s s

ptpdc

block IGBT ac NN

V

V V

module,

,

,

1

2 















+= (Ref. 112) 

With Vac as the peak nominal line-to-neutral AC side voltage, Vdc,ptp as the pole-to-pole nominal DC side 

voltage, Ns,module as the number of series IGBTs within the module blocking the voltage (1 for half bridge, 

2 for full bridge), and Ns as the number of series IGBT modules. Ns is calculated so that Vblock,IGBT is 

within the IGBT scaling data and for the lowest number of IGBTs in order to minimize the IGBT and 

total converter mass. 
NASA/CR—2015-218440 60

The steady-state rated conduction current through the IGBT module is: 

p

acdc

IGBT N

II I 1

23 



 



 += (Ref. 113) 

With Iac as the peak nominal AC side current, Idc as the nominal DC side current, and Np as the number of 

parallel IGBT modules. Within the module, there are no parallel-connected IGBT modules. Np is 

calculated so that IIGBT is within the IGBT scaling data (below the maximum current but not necessarily 

above the minimum current) and for the lowest number of IGBTs. 

Trends from this existing IGBT data follow and are used to estimate a scalable IGBT mass and losses 

for given nominal current and blocking voltage ratings (Figure 41 and Figure 42). The turn-on time used 

for scaling is the average of the IGBT turn-on time data (2.128 μs) since it did not vary significantly with 

rated voltage or current. 

The IGBT mass in kg is estimated by the trend as a function of nominal current in A and nominal 

blocking voltage in kV by the equation: 

( ) ( )

( ) ( ) ( )2 , , ,

, , kg,

871.1

215.4

0102.0

871.1

215.4

6.376

6.884

0413.0

871.1

215.4

173.0

6.376

6.884

350.0405.1

− + − − +

− + − +=

C rated block rated block rated

C rated block rated

IGBT

I V V

I V

Mass

Note that this IGBT mass data includes the antiparallel free-wheeling diode in the same module. For the 

current-source converter topology, the diode is in series with the IGBT rather than free-wheeling. For 

similar voltage and current ratings as the IGBT, the diode mass estimate is included as part of the IGBT 

mass estimate. 

The IGBT turn-off energy in J is estimated by the trend as a function of nominal current in A and 

nominal blocking voltage in kV by the equation: 

( ) ( )

( ) ( ) ( )

( )2 ,

, , 2 ,

, , ,

871.1

215.4

1364.0

871.1

215.4

6.376

6.884

801.0

6.376

6.884

101.0

871.1

215.4

54.1

6.376

6.884

8858.0319.2

− −

− − + − +

− + − +=

block rated

C rated C rated block rated

C rated block rated

off IGBT

V

I I V

I V

E

Once the individual IGBT and diode mass and loss estimates are calculated for the specified current and 

voltage ratings, the total IGBT and diode mass and losses are calculated based on the converter topology. 

The IGBT and diode total mass for the rectifier/inverter CSC topology is calculated by 

MassIGBT,total = IGBT NNNMass AMpAM

Where NM is the number of IGBTs per switching module (2 for half bridge, 4 for full bridge), NMpA is the 

number of switching modules per arm based on the series/parallel configuration, and NA is the number of 

arms for the converter (6 for the midpoint clamped topology). This same equation is used to calculate the 

IGBT and diode mass since each IGBT is paired with a series diode and the IGBT mass estimate includes 

the diode mass estimate (Figure 43). 
NASA/CR—2015-218440 61

Figure 41.—IGBT Conduction Voltage Drop Data 

and Trends versus Blocking Voltage for Room 

Temperature IGBT Scaling.

Figure 42.—IGBT Turn-Off Time Data and 

Trends versus Nominal Current for Room 

Temperature IGBT Scaling.

The total loss contribution of the IGBTs and diodes is described by the equation: 

( )

( ) AMpA

M

cond IGBT sw IGBT diodecond sw diode

AMpA

M

loss Subtotal loss IGBT diodeloss

NN N P P P P

NN N P P P

2

2 2

2

, , , ,

, , ,

= + + +

= +

Where Pcond and Psw are the conduction and switching losses, respectively. These power loss calculations 

are described in the next few pages. The number of IGBTs or diodes per module is halved for the 

conduction loss contribution since only half of the devices are conducting at any time. 

The overall converter model specifies the series/parallel configuration of the converter IGBTs so that 

each device has voltage and current ratings within this range (if possible) so as to minimize the 

extrapolation of the scalable IGBT trends. It will be shown how extrapolation of these trends affects the 

high power rectifier total mass and losses. In order to avoid extrapolation and potentially achieve 

smoother total converter trends, additional existing IGBT data can be included in the table, and revised 

trends can be developed. Additional data for lower current rated IGBTs is necessary to avoid 

extrapolation of the existing trends for lower power devices rated for higher DC voltages. An alternative 

approach, as cryogenic device and converter operation is further researched and developed, would be to 

develop IGBT models based on required material and configuration in order to achieve the desired 

voltage, current, and interruption time ratings. Such a model requires in-depth knowledge of the device 

physics and material science. An example of the construction of punch-through and non-punch through 

IGBTs is shown in Figure 44. 

To estimate the power losses of the IGBT and diode devices, the switching and conduction losses are 

estimated and summed. The conduction and switching losses for standard voltage-source two-level 

inverters and rectifiers are well known and can be used to approximate the conduction and switching 

losses for this converter topology for this system study. An improved estimate of the losses can be 

calculated using circuit simulations. 
NASA/CR—2015-218440 62

For the use of IGBTs and diodes in an inverter or rectifier, the converter is operated using pulse-width 

modulation (PWM). The IGBT conduction losses for a PWM-controlled converter are approximated by: 





 





π

+φ+















+

π = 2

2

, 3

1

8

cos

2 4

1

CE C

C

CE

C

CE

C

cond PWM CE Ir I Vm

I

r

I P V (Ref. 114) (3) 

Where VCE is the collector-emitter voltage during conduction, IC is the collector current, rCE is the 

equivalent collector-emitter resistance, and m cos φ is the equivalent PWM duty cycle for a given 

operating point. The value used for m cos φ to compute rectifier operation conduction losses throughout 

this study is 64.0 2 −≈ π

− . For inverter operation, the value used is 0.64. Nominal conduction rCE can be 

computed for rated conduction VCE and IC by:

C

CE

CE I

V r =

The diode conduction losses may be calculated using the same for IGBT conduction losses since they are 

in series for the current-source configuration rather than antiparallel for the traditional voltage-source 

configuration. In that case, VCE is instead the diode forward voltage, IC is the diode conduction current, 

and rCE is the diode on-state resistance.

Figure 43.—IGBT Turn-On Energy Data and Trends 

versus Blocking Voltage for Room Temperature IGBT 

Scaling. 
NASA/CR—2015-218440 63

Figure 44.—PT and NPT IGBT Structure (Ref. 115). 

IGBT or diode switching losses for a PWM-controlled converter are approximated by: 

( ) Psw PWM sw + EEf offon π = 1 , (4) 

Where fsw is the converter switching frequency, Eon is the turn on energy, and Eoff is the turn off energy. 

When IGBTs and diodes are used in a DC/DC converter, such as the bidirectional converter used to 

interface the SMES to the DC distribution bus, the converter duty cycle (D) drives switching losses. The 

IGBT or diode conduction losses for use in a chopper configuration (DC/DC converter) are approximated 

by:

( )DIrIVPcond CE C CE C

2 += (5)

With each switch turning on and off once per cycle, the switching losses for the DC/DC converter are 

approximated by: 

swsw ( ) += EEfP offon (6) 

5.2.2.2.3 Capacitor Sizing Model

The specified capacitance, voltage, and current from the overall converter model are used to estimate 

the capacitor mass and losses. The total converter capacitance requirement is determined by: 

dcac

dc

eq Vf

I C

∆ = 2
NASA/CR—2015-218440 64

For the converter topologies modeled, the equivalent capacitance is divided among six separate 

capacitors, one for each arm. For three identical capacitors in parallel, all in series with an additional set 

of three identical capacitors in parallel, the individual arm capacitance is calculated by 

arm CC eq 2

3 =

The average arm capacitor voltage is: 

2

, ptpdc

Carm

V

V =

The estimated arm capacitor current is the total arm current. However, the arm capacitor current may even 

higher depending on the DC bus voltage ripple and frequency. 





 



 +≈

23

acdc

Carm

II I

These arm capacitor specifications are given to the capacitor sizing model to estimate the arm 

capacitor mass and equivalent series resistance (ESR). A cylindrical film-foil type capacitor is used to 

model the arm capacitor. This type of capacitor is chosen because the permittivity of the dielectric, 

polypropylene, used in high power capacitors performed well when tested in cryogenic environments 

(Refs. 116 and 117). When tested at 77 K, 0.99 μF room temperature rated capacitors had the same 

capacitance at 77 K, and the ESR decreased by more than a factor of 2. Film capacitors are widely used as 

DC link capacitors and IGBT snubbers such as the capacitors for these power converters. Aluminum was 

chosen as the foil electrode material for this capacitor scaling (Ref. 118). A different metal may be used 

to further reduce the capacitor ESR. An example of a metalized polypropylene film capacitor construction 

is shown in Figure 45. 

To construct the capacitor, layers of foil and dielectric film are added in cylindrical form until the 

capacitor has the desired capacitance rating. The capacitance of each foil+dielectric+foil set of layers is 

estimated by the capacitance of a cylinder:





 



 επε=

a

b

L Clayer r

ln

2 0

Where εr is the relative permeability of the dielectric, ε0 is the permittivity of a vacuum (constant at 

8.8541878176 * 10–12), L is the cylinder length, b is the outer conductor radius, and a is the inner 

conductor radius. This equation is valid when L is large compared to b. For polypropylene dielectric, εr is

2.2-2.36. The value used for this modeling is 2.2. A fixed length of 0.1 m is used for the capacitor models 

throughout this study. The capacitor model could be modified to minimize mass while allowing the length 

and other parameters to vary. 
NASA/CR—2015-218440 65

Figure 45.—Metallized Polypropylene Film Capacitor Construction from ABB (Ref. 119). 

To estimate the outer and inner conductor radii, material properties for polypropylene film capacitors 

are used. The standard film thickness for polypropylene is 2.4 to 3.0 μm, and its dielectric strength εs is 

650 V/μm. The modeled film thickness in μm is described by: 

















ε = s

arm t

V F Maximum 2.4,of

so that the dielectric thickness meets the minimum capacitor voltage rating. This allows only one 

capacitor to be modeled for a given capacitance and voltage rating. The foil thickness Mt is assumed to be 

twice the dielectric thickness. These dielectric and foil thicknesses are the same for every cylindrical layer 

used to form the capacitor. 

From these thicknesses and the cylindrical capacitance estimate equation, the number of film foil 

layers Nlayers is calculated to form the capacitor with the specified capacitance by adding a layer, 

computing the layer’s capacitance, and checking whether the total capacitance thus far is greater or equal 

to the specified capacitance. If it is less, then another layer is added. 

Once the number of film foil layers is determined, the cylinder radius is calculated by: 

( ) cyl = layers + MFNr tt
NASA/CR—2015-218440 66

The mass of the cylinder is estimated by calculating the mass of the dielectric and metal layers from the 

dielectric and metal volume and density. Constants used are the polypropylene density of 946 kg.m3

 and 

Aluminum density of 2700 kg/m3

The capacitor ESR is calculated by the equation:

( ( ) ) 2 2

rr Mt

LM ESR

cyl cyl

r

−−π

=

where the Mr is the metal resistivity (Aluminum used for the conductor has a resistivity of 28.2 nΩ-m).

This is the resistance for the outer conductor layer. 

The room temperature capacitor scaling model results were checked against Cornell Dubilier 

polypropylene round axial film capacitor ratings and masses (Ref. 120) to understand the accuracy of this 

capacitor scaling method. 

The cryogenic scaling of this capacitor is based on the test results of polypropylene film capacitors at 

77 K (Refs. 121 and 122). Relevant results from this study are summarized in Table 16. Based on this 

results, the room temperature capacitor is scaled to a cryogenic environment by a 1:1 ratio for 

capacitance, and multiplying the room temperature ESR by 0.5. Also, a packing material improvement 

factor for the N+3 application is used. In this case, it is specified as 0.7, which indicates the capacitor 

mass can for N+3 is approximated as 70 percent of the present capacitor mass. 

5.2.2.2.4 Inductor Sizing Model

Given the total lumped arm inductance from Equation (2), the individual switching module 

inductance specifications are determined. From those specifications, the mass and loss of each inductor is 

estimated by the inductor model which is based on the SFCL model. For detail on the SFCL model, see 

Section 5.6. 

The switching module link inductance is calculated by:

arm

s

p L

N

N

Lmodule =

TABLE 16.—CRYOGENIC 100 VDC POLYPROPYLENE CAPACITOR OPERATION AT 77 K

Capacitor 

rating

Control at room temp In-situ LN2 Conclusions

Capacitance, 

μF

Frequency,

kHz

Dissipation 

factor x10–2 

ESR, 

Ω

Capacitance, 

μF

Dissipation 

factor x10–2 

ESR, 

Ω

ESR ratio 

cryogen/room 

temperature

Capacitance 

ratio cryogen/

room 

temperature

0.99 1 0.02 0.03215 1 0.01 0.01591 0.495 1.01

0.99 20 0.18 0.01445 1 0.08 0.00637 0.44 1.01

0.99 50 0.49 0.01575 1 0.31 0.00987 0.626 1.01
NASA/CR—2015-218440 67

The module inductor current rating is the nominal AC peak current. The rated voltage is 

approximated by: 

( ) acacdc

s

L IfI N

L

dt

di V = L ≈ 2π+∆ 1

module

module

module module

These specified inductance, voltage, and current ratings are then used by the inductor model to estimate 

the module inductor mass and losses. The main difference between the inductor model and SFCL model 

is that there is no quench state resistance for which the inductor is sized; the inductor is sized to meet the 

desired inductance only. 

5.2.2.2.5 Packaging Model 

The packaging mass estimates, which include the heat exchanger, housing, bus bars, and current sensors, 

are based on scaling of state-of-the-art power converters. This was discussed previously in Section 3.5. 

Using the data and studies from references (Refs. 123 and 124), mass percentages of the packaging 

components were estimated and are listed in Table 17. Using these percentages, the packing component 

masses are scaled according to the subtotal mass of the inductor, presspack diodes, and IGBT and series 

diodes as estimated by the model. More detailed models of each of these components could be described 

so that they are more independent of the IGBT and inductor scaling models. Some ways in which these 

models could be developed and scaled are: 

• The heat exchanger mass could be estimated based on the converter losses and thermal transfer 

capability. 

• The housing could be based on a volume estimate of the converter. 

• The bus bars could be estimated by the rated AC and DC currents and estimated connections of 

all of the circuit components. 

• The current sensors could be scaled as a function of the AC and DC currents.

For such packaging models, these models should be verified against the existing state-of-the-art high 

power electronic devices. For the N+3 time frame and this sensitivity study, using scaling to estimate the 

packaging mass is assumed to be appropriate. 

5.2.3 Source-Side Converters Trends

The unidirectional rectifiers are used as the power converters to convert the generator AC voltage to DC 

bus voltage. With four generators for this architecture, these rectifiers are rated for 50 percent of the 

minimum takeoff power of 25 MW. This power rating is 12.5 MW. The parameters used to develop the 

following trends are described in Table 17. 
NASA/CR—2015-218440 68

TABLE 17.—ASSUMED PARAMETERS AND VALUES FOR

UNIDIRECTIONAL CURRENT-SOURCE CONVERTER TRENDS

Parameter Value

Converter 

Sizing

Modulation index................................................................................................................................. 0.8

Power factor............................................................................................................................................1 

Operating temperature ....................................................................................................................100 K 

DC voltage ripple magnitude, % ...........................................................................................................20

DC current ripple magnitude, % ...........................................................................................................10

Inductor

DC pole-to-pole resistance for inductance requirement......................................................................1 Ω

Insulator dielectric strength (LPP).........................................................................................100 kV/mm

Ambient temperature of the superconductor and cooling reservoir ................................................100 K 

Superconductor critical temperature ..............................................................................................110 K 

Flux-creep region exponent (at 77 K) ....................................................................................................6 

Flux-flow region exponent .....................................................................................................................3 

Initial critical current density (at 77 K), i.e., current density where E=1μV/cm ................ 1.5×108 A/m2

Initial electric field.................................................................................................................0.0001 V/m

Electric field at transition from flux-creep state to flux-flow state .............................................0.1 V/m

Coefficient for heat transfer to cooling reservoir................................................................1500 W/K m2

Superconductor density.......................................................................................................... 6500 kg/m3

Dielectric insulation density .................................................................................................... 900 kg/m3

Former density....................................................................................................................... 1800 kg/m3

Coolant medium................................................................................................................................... N2

Former thickness............................................................................................................................. 2 mm

Inductor L/D limit................................................................................................................................. 10

IGBT and 

Diode

N+3 mass improvement scaling factor................................................................................................. 0.9

Ideal duty cycle....................................................................................................................................0.5

Conduction modulation (PWM)....................................................................................................... –0.64

Capacitor

Cylinder length ................................................................................................................................0.1 m 

Capacitor scaling cryo/room temperature ...............................................................................................1 

ESR scaling cryo/room temperature ....................................................................................................0.5

Improved materials and packaging = (N+3 mass)/(N mass)................................................................0.7

From Figure 46, the nonlinear specific power trend for the high power bidirectional CSC versus DC 

bus voltage is shown. Note that the discontinuities in these trends occur because of the switching module 

series/parallel configuration changes as the DC bus voltage increases. As described previously, the IGBTs 

are scaled for a certain current and voltage range. To interpolate the IGBT data, a series/parallel 

configuration of the IGBTs is required for operation of the converter beyond the current and voltage for 

which IGBT data is scaled. The series/parallel configuration is graphed in Figure 47 and should be used to 

understand the discontinuities. 

For each of the AC frequency trends, the specific power is maximized for a DC bus voltage of 

5.5 kV. Looking at one AC frequency trend, the below the 5.5 kV DC bus voltage, the specific power 

increases (mass decreases for fixed power). Above the 5.5 kV DC bus voltage, the specific power 

gradually decreases (mass gradually increases). This trend can be further explained by analyzing the 

converter mass breakdown by component. Figure 48 shows the converter mass breakdown for a 100 Hz 

AC system, and Figure 49 shows the converter mass breakdown for an 800 Hz AC system. 

For the lower bus voltages, the capacitor mass a high percentage of the total converter mass. For the 

higher bus voltages, the inductor mass increases but is not as significant a portion of the total converter 
NASA/CR—2015-218440 69

mass as is the capacitor. This mass breakdown shows that the high power unidirectional power converter 

mass is sensitive to the DC bus voltage ripple and capacitance required at lower DC bus voltages. This 

trade-off between capacitance and inductance requirements and DC bus voltage is shown in Figure 50. 

Comparing Figure 48 and Figure 49 for lower and higher AC frequency systems, respectively, the 

trends clearly indicate a decreased total converter mass as well as decreased capacitor and inductor mass 

for the higher frequency AC system. These mass breakdowns show that the high power bidirectional 

converter mass is sensitive to the AC system frequency. The converter structure mass (current sensors, 

bus bars, housing, heat exchanger) are scaled with IGBT + diode + inductor mass. 

The total IGBT and diode mass increases as the DC bus voltage increases. This can be attributed to 

the IGBT mass scaling as a function of voltage and current and required number of series IGBTs to block 

the required voltage. 

Figure 46.—Unidirectional CSC Specific Power versus DC Voltage for 

Several AC Frequencies. 

Figure 47.—Unidirectional CSC Number of Parallel and Series 

Switching Modules versus DC Voltage. 
NASA/CR—2015-218440 70

Figure 48.—Unidirectional Converter Mass Breakdown for 

100 Hz AC Frequency. 

Figure 49.—Unidirectional Converter Mass Breakdown for 

800 Hz AC Frequency. 
NASA/CR—2015-218440 71

Figure 50.—Unidirectional CSC Inductance and Capacitance. 

Figure 51.—Unidirectional CSC Efficiency versus DC Voltage for Several 

Switching Frequencies. 

Figure 52.—Unidirectional CSC Losses versus DC Voltage for Several 

Switching Frequencies. 

Figure 51 and Figure 52 show the trends of the converter efficiency and losses for several switching 

frequencies. As the DC bus voltage increases, the efficiency decreases. This decrease is more pronounced 

for higher switching frequencies. Also, higher switching frequencies correspond to the decreased overall 

efficiency. This can be further explained by examining the switching and conduction losses that form the 

total power loss. An example of these trends for a 5 kHz switching frequency is shown in Figure 53. As 

the DC bus voltage increases, the conduction losses decrease due to the lower conduction current.

However, the switching losses increase due to the scaling of the IGBT turn-on and turn-off energies. 
NASA/CR—2015-218440 72

Figure 53.—Unidirectional CSC IGBT and Diode Total Switching 

and Conduction Losses versus DC Bus Voltage for an Example 

400 Hz AC System, 5 kHz Switching Frequency. 

Figure 54.—Unidirectional CSC Interruption Time versus DC Voltage. 

Figure 54 shows the trend of the converter interruption time versus DC distribution voltage.

Generally, the interruption time decreases with increased DC voltage, which may be advantageous for the 

protection system in order to limit the amount of fault current through the system. The discontinuities in 

this trend are attributed to the change in parallel switching module configuration. The interruption time is 

based on the IGBT turn off time which is scaled as a function of current. IGBTs are arranged in parallel 

for total current ratings exceeding the scaling range. 

Figure 55 combines several of these model trends to assess the converter losses as a function of 

interruption time. This trend indicates that to achieve interruption times less than 1 μs, the converter will 

have additional losses. For interruption times greater than 1 μs, the converter losses are approximately the 

same. 
NASA/CR—2015-218440 73

Figure 55.—Unidirectional CSC Losses versus Interruption Time for Several 

Switching Frequencies. 

5.2.4 Load-Side Converters

As previously discussed, the load-side converters require bidirectional current flow. In order to 

achieve this, a similar topology is used, but the switching modules must be bidirectional. Such switching 

modules are formed in the full bridge configuration. 

Current source converters may be used for the load side as well as the source side as long as only one 

of the converters controls the current and there is one or more “sink” buses through which energy not 

used can be stored or additional energy required can be fed to the system. These “sink” buses are the 

SMES units located on each of the four distribution buses. 

The topology chosen for the bidirectional CSC is similar to that for the unidirectional CSC with the 

exception of the switching modules. Figure 56 illustrates the bidirectional CSC topology, and Figure 57 

illustrates several of the switching module circuit options. These switching module options are similar to 

those for the unidirectional CSC except that they are arranged in a full bridge configuration to enable 

bidirectional current flow. Note that as with the unidirectional switch modules, only half of the IGBTs 

conduct current at any given time. For the full bridge case, that means that two IGBTs and series diodes 

conduct at a time. Thus the conduction losses include two IGBT and series diode conduction losses. The 

individual IGBT voltage rating is half the overall switching module rating but must carry the full 

switching module current. 

The switch module topology options shown in Figure 57 include a full bridge without overvoltage 

protection (Figure 50(a)), full bridge with thyristors for overvoltage protection (Figure 57(b)), and full 

bridge with presspack diodes for overvoltage protection (Figure 57(c)) (Ref. 91). The topology in 

Figure 57(c) was chosen for this model. The converter components are all scaled the same way as for 

the unidirectional CSC but have voltage and current ratings for the switching module topology in 

Figure 57(c). Note that, similar to the unidirectional converter model, the required capacitance is lumped 

as a single arm capacitor rather than distributed to each switching module. It is possible that the 

simulation or implementation of this converter topology may indicate that a capacitor is required for each 

switching module, rather than a lumped capacitor, in order to equally divide the voltage among each of 

the series-connected switch modules.
NASA/CR—2015-218440 74

Figure 56.—Bidirectional, Current-Source Converter Topology 

Selected for the Propulsor Motor Drive. The Yellow Blocks in this 

Diagram Represent Potential Variations in the Switch Topologies.

Figure 57.—Full-Bridge, Bidirectional, Current-Source Converter Semiconductor Switches. Topology 

C (Presspack Diodes for Overvoltage Protection) Were Selected for the Switch Topology for the 

AC to DC Converter over Topology B (Thyristor for Overvoltage Protection). 
NASA/CR—2015-218440 75 

5.2.5 Load-Side Converter Trends

The bidirectional converters are used as the motor drives and are all rated for 7.14 percent of the 

minimum takeoff power 25 MW. This power rating is 2.5 MW. The assumed parameters for which these 

trends were developed are described in Table 17 and in Section 5.2.2.2 except that the conduction 

modulation for PWM is positive 0.64 for inverter operation. 

When interpreting these trends, it is important to note that development of these trends required 

extrapolation of the IGBT trends. The IGBT trends were based on room temperature IGBT current ratings 

200 to 1500 A, but for this lower power rated converter, the room-temperature translated IGBT current 

rating is less than 200 A when the DC pole-to-pole converter bus is greater than 4.5 kV. Consequently, 

for system-level sensitivity studies at higher voltages, existing IGBT data for lower current levels but 

similar voltage levels need to be added to the IGBT scaling for higher confidence in the converter model 

trends.

From Figure 58, the specific mass versus voltage does indicate a nonlinear trend at which the 

bidirectional converter’s mass can be minimized (specific mass maximized for the given power rating). 

For each AC frequency, the maximum specific power occurs at 2.5 kV. With increasing AC frequency, 

the specific power trend is higher. This is due to the smaller inductance and capacitance required for 

higher AC frequencies for fixed DC voltage and current ripple magnitudes. The smaller inductance and 

capacitance requirements generally result in lighter devices. 

The general trend for specific pass can be explained by analyzing the converter mass breakdown in 

Figure 59 and Figure 60. 

For lower DC voltages, the capacitor mass is larger and is a large percentage of the overall converter 

mass. For higher DC voltages, the capacitor mass decreases while the inductor mass increases. The 

percentage of the capacitor and inductor mass relative to the total converter mass is due to the scaling of 

the packaging components (heat exchanger, hosing, bus bars, and current sensors) relative to the baseline 

mass breakdown and the estimated IGBT and series diode total mass. 

Figure 58.—Bidirectional CSC Specific Power versus DC Voltage for Several AC Frequencies.
NASA/CR—2015-218440 76

Figure 59.—Bidirectional Converter Mass Breakdown for 100 Hz AC Frequency. 

Figure 60.—Bidirectional Converter Mass Breakdown for 800 Hz AC Frequency. 
NASA/CR—2015-218440 77

The capacitor mass is larger for lower DC voltages due to the capacitance requirements and the 

relationship between the rated DC current and DC ripple voltage magnitude from Equation (1). The 

decrease in rated DC current for the fixed converter power at higher DC voltages drives the decrease in 

capacitance for the fixed DC bus voltage ripple. Generally, as explained in the capacitor model in 

Section 5.2.2.2.3, the higher the capacitance required, the more film-foil layers required for a fixed 

capacitor length, and the heavier the capacitor. The increase in inductor mass for higher voltages is 

generally due to the increase in individual and total inductance requirement which requires more mass. 

The discontinuous changes in the inductor mass are due to the addition of a series switching module as 

required for higher voltages and the need to interpolate IGBT data for scaling purposes. 

Figure 61 shows the equivalent capacitance and arm inductance requirements versus DC bus voltage 

for a 400 Hz AC frequency. 

Figure 62 and Figure 63 show the CSC efficiency and loss trends. Generally, for a specific switching 

frequency, as the DC bus voltage increases, this converter model at the specified power level indicates an 

increase in losses As the DC bus voltage increases, the individual switching module current rating 

decreases, and this decreases the conduction losses. However, the turn-on energy increases according to 

the IGBT scaling trends, and this increases the switching losses. Note that the turn-on energy is scaled as 

a function of IGBT rated voltage. Figure 64 illustrates an example trend of the IGBT and diode switching 

and conduction losses versus DC bus voltage. 

Figure 61.—Bidirectional CSC Inductance and Capacitance Requirements versus DC 

Bus Voltage for 400 Hz AC. 
NASA/CR—2015-218440 78

Figure 62.—Bidirectional CSC Efficiency versus DC Voltage for Several Switching 

Frequencies. 

Figure 63.—Bidirectional CSC Losses versus DC Voltage for Several Switching 

Frequencies. 

Figure 64.—Bidirectional CSC IGBT and Diode Total Switching and 

Conduction Losses versus DC Bus Voltage for an Example 400 Hz

AC System, 5 kHz Switching Frequency. 

For a higher switching frequency, the switching losses increase due to more switching events per 

second, and the total converter losses increase. For this converter power rating, the switching losses are 

more significant than the conduction losses even at lower DC bus voltages. 

Figure 65 illustrates the series/parallel configuration of the switching modules for each bus voltage. 

For this low power rating, only one module is required (none in parallel) due to the lower current rating 

and IGBT current scaling range. As the DC bus voltage range increases, more switching modules are 
NASA/CR—2015-218440 79

configured in series to divide the pole-to-pole DC bus voltage. This trend is provided so as to help the 

reader understand the locations of the discontinuities in the trends. 

Figure 66 indicates a decrease in interruption time for higher rated DC bus voltage. This is due to the 

IGBT scaling trend for turn-off time as a function of rated current as shown in Figure 42. Note that the 

turn-off time trend is extrapolated for DC pole-to-pole converter bus voltage greater than 4.5 kV. Further 

confidence in these turn-off times can be gained with additional lower current rated IGBT data and trend 

development. 

Figure 67 shows a combination of the trends presented with total CSC losses versus interruption time. 

This shows that, based on the model data, there is a trade-off in lower losses and faster interruption time 

especially for the interruption times less than 1 μs. For longer interruption times, lower losses generally 

result from lower switching frequencies. 

Figure 65.—Bidirectional CSC Number of Parallel and Series Switching Modules versus

DC Voltage. 

Figure 66.—Bidirectional CSC Interruption Time versus DC Voltage (Extrapolated IGBT 

Data).

Figure 67.—Bidirectional CSC Losses versus Interruption Time for Several 

Switching Frequencies (Extrapolated IGBT Data). 
NASA/CR—2015-218440 80

5.3 Cables Model Overview

The cable models determine the mass efficiency and losses associated with the previous estimates 

determined that the cable system contributes roughly 4 percent of the overall mass of the TeDP microgrid 

system. However, the overall contribution of cables to system weight depends on the operating voltage 

(ACRMS or DC), the power rating for the cable, and its length. The cable model was defined following the 

general parameter structure illustrated in Figure 68. 

As illustrated in this figure, the outputs of interest are cable weight and efficiency, as well as the loss 

properties. This includes the heat loss at the rejection temperature (heat quality) as well as the power 

required for coolant pumping. The cable design is sensitive to the fault current margins required, the 

topology of the cable, the superconductor type, and the operating temperature of the cable. 

Cable impedance values are also desired for the cables. This information will be used to determine the 

fault response for the architecture. Current impedance information reported from the cable model is 

limited to estimated resistive losses based on cable length. 

5.3.1 Cable Layout 

There are many superconducting cable concepts (Ref. 125). Generally, cable concepts fall into two 

categories: coaxially arranged cables with multiple conductors and parallel runs of single conductor 

cables. For the purposes of this model a Nexans Triax cable configuration was selected (Ref. 126). 

Figure 68.—Parameter Diagram for Cable Model. 
NASA/CR—2015-218440 81

Figure 69.—Modeled Cable Layout. 

As illustrated in Figure 69, this cable formed around a hollow mandrel through which liquid coolant 

is pumped. This mandrel is wound with layers of superconducting material and dielectric insulation. 

These layers are bathed in another layer of coolant. This coolant is contained within a vacuum cryostat 

lined on the inner surface with layered Mylar film insulation to reduce radiated heat from the outside 

environment to the cable. 

The Nexans configuration includes a copper EMI screen for three phase AC applications. For the bipolar DC configuration of this cable the copper screen is removed. 

The lack of quench conductors in this Nexans cable requires that the superconductor be sized to 

withstand all overcurrents for the systems. Therefore, the protection systems must be configured in a 

manner which allows enables this capability. Including superconducting fault current limiters in the 

protection system provides support for this assumption. With adequate protection, quench does not occur 

in the cable itself, but is managed at the fault current limiter pinch points. 

Four cable configurations are modelled with this cable topology. Three of these calculations DC: 

Monopolar, Bipolar with 100 percent single pole loss capability, and bipolar with 50 percent single pole 

loss capability. The final configuration is a three phase AC capability. Both bipolar DC configuration and 

the three-phase AC configuration take advantage of the Triax cable layout, while the monopolar DC 

configures only requires a two wire coaxial layout.

5.3.2 Monopolar, Bipolar, Redundancy and Three Phase AC

Sizing for these different configurations is driven primarily by the carrying current for each 

conducting element and the strength of the dielectric material required between each superconducting 

layer. For the three-phase AC cable, the insulation is configured to manage the max phase to phase 

voltage differential. The bipolar configuration requires a dielectric thickness depending on the voltage 

differential applied between the central ground conductors during a single phase short scenario. In order 

to minimize dielectric thickness, the neutral or ground conductor is situated between the two positive and 

negative conductors in the bipolar configuration.
NASA/CR—2015-218440 82

5.3.3 Superconducting Material

The required thickness of the superconducting material is based on its current carrying capacity. The 

critical current density (Jc) for a superconducting material depends is a function of the operating 

temperature the magnetic to which the material is exposed. A comparison of superconducting 

performance for various superconducting material is Figure 70. 

Specific trends of critical current density for 3 superconducting material were included in the cable 

models. These materials are YBCO, MgB2, and BSCCO. The applied field and operating temperature 

scaling of critical current density of these materials were applied according to the relationships shown in 

Figure 71 and Figure 72. 

Figure 71 illustrates the sensitivity of YBCO tape to operating temperature and both a parallel and 

perpendicular magnetic field. YBCO exhibits preferable superconducting capability. Especially at very 

low temperatures, the critical current density is very high and insensitive to the applied magnetic field. 

The published critical current densities for HyperTech’s BSCCO round wire are illustrated in 

Figure 72. This graph gives the material critical current density (Jc) and the engineering critical current 

density (Je). The engineering critical current density is given as roughly an order of magnitude below the 

material critical current density. This scaling of performance is implemented in the cable sizing model via 

a performance factor of 0.1 applied to Jc. 

Figure 70.—Comparison of Critical Current Density for Various Superconducting Materials (Ref. 127). 
NASA/CR—2015-218440 83

Following the application of the Nexans cable concept, it is assumed that the cable applies a wrapped 

tape construction approach. The cross sectional area of the conductor is determined by assigning a max 

operating current and estimating the critical current density based on temperature and field. This then 

determines the thickness of the tape required.

Conductor mass depends on the material applied. Density values of 6.38, 2.57, and 6.5 g/cm3 were 

applied for YBCO, MgB2, and BSCCO, respectively. 

5.3.4 Applied Field

For the purposes of the cable sizing it was assumed that there is no externally applied field acting on 

the cables. 

5.3.5 Cryostat 

The cryostat performance and weight approximations were estimated using published geometry, 

performance, and weight data from Nexans Cryoflex cryostat datasheets (Refs. 128 and 129). Cryoflex 

consists of two coaxially configured corrugated metal tubes, the inner wrapped with layered Mylar 

insulation. A vacuum is pulled between these layers to minimize the heat leakage into the cable from the 

environment. The layout and cable information are given in Figure 73. 

Figure 71.—YBCO Critical Current Density Sensitivity to 

Temperature and Field (Ref. 130). 
NASA/CR—2015-218440 84

Figure 72.—MgB2 Critical Current Density Sensitivity to Temperature and Field (Ref. 131). 

Figure 73.—Cryoflex Physical Properties (Ref. 132). 

Trends for cryostat outer diameter as well as the overall insulation system mass were created given 

this data and are illustrated in Figure 74 and Figure 75. These trends were used to estimate the physical 

properties of the cable system.

It was assumed that the inner cryogenic return fluid flow tube around which the superconducting 

material is wound has a diameter of 5 mm. The cross-sectional area of the outer coolant flow channel is 

the same as the inner. 

The wall thickness of the corrugated stainless steel tubing was assumed to 1 mm and the roughness 

coefficient was assumed to be 0.02 for Reynolds number flow calculations 
NASA/CR—2015-218440 85

Figure 74.—Trend for Cryostat Outer Diameter in as a Function 

of Cryostat Inner Diameter from Cryoflex Data.

Figure 75.—Trend for Cryostat Linear Mass as a Function of 

Inner Diameter from Cryoflex Data.

TABLE 18.—COOLANT PHYSICAL PARAMETERS ASSUMPTIONS

LN2 LH2

Density, kg/m3 810 71

Dynamic viscosity, μPa 140 150

Specific heat 11.8 at 20 K 2.23 at 70 K 

5.3.6 Cooling 

Data from both liquid hydrogen and liquid nitrogen was used in this model depending on the type of 

superconductor selected. Coolant parameters are given in Table 18. 

5.3.7 Losses

The role of the cryostat is to maintain appropriate operating temperature for the superconducting cables. 

The primary source of inefficiencies for a DC superconducting cable system is not due to losses in the cable 

itself, but heat leakages into the system from the environment (Refs. 133 and 134) As such, the heat leakage 

into the cable was estimated using the Cryoflex physical data (Ref. 135), illustrated in Figure 76. 
NASA/CR—2015-218440 86

Figure 76.—Heat Leakage per Meter from the Environment to the 

Cryofluid from Cryoflex Data.

Additionally, research shows a potential 34 percent improvement in cryostat performance is available 

with alternative cryostat configurations (Ref. 136). Therefore, this estimated heat leakage value can be 

scaled to represent future technology improvements.

5.3.8 AC Losses

AC superconducting cables introduce additional conduction losses which effect coolant flow 

requirements (Ref. 137). The AC losses are a log-log function of the RMS current and the AC frequency. 

Figure 77 illustrates the trend used to estimate the losses per unit length for the AC portions of this 

superconducting microgrid. 

The losses on a superconducting AC cable are a function of frequency and the ratio between the 

critical current and the operating current. In this report, this is termed the critical current ratio. The critical 

current ratio is used in the cable to trade between conduction losses and additional conductor mass. 

DC cable efficiency is also a function of critical current ratio. This trend is illustrated in Figure 78. 

5.3.9 Dielectric

The dielectric referenced in this study was laminated polypropylene paper (LPP) due to its favorable 

performance at low temperature. Referencing Cheon et al., (Ref. 138), it was assumed that the dielectric 

strength was 50 kV/mm. While this value is sensitive to the number of layers of the LPP and the electrical 

signal (AC, DC, or Impulse) and the pressure, this assumption provides a reasonable estimate for 

dielectric performance. Dielectric performance is illustrated in Figure 79. 

The thickness and weight of the LPP depends on the off-stand voltage and the material density. For 

the 100 percent redundant bipolar configuration, each layer of dielectric insulation must be able to 

withstand the nominal off-stand voltage. During a failure of the negative pole, the neutral conductor 

operates at the negative voltage. For the 50 percent redundant system the dielectric material must 

withstand 50 percent of the total off-stand voltage; maintaining the positive or negative pole with respect 

to a neutral ground conductor. During a failure of the negative pole in this scenario, the neutral pole 

remains neutral and the power available is reduced to half to the original capability.

For the AC case, the dielectric is sized considering the peak to peak voltage. The density of the LPP 

was set at 900 kg/m3 (Refs. 139 and 140). 
NASA/CR—2015-218440 87

Figure 77.—Superconductor AC Losses (Ref. 141). 

Figure 78.—Superconducting DC Losses (Ref. 141). 

5.3.10 Mass and Efficiency Trends

This section presents the sensitivity of the cable mass, diameter, and losses in terms of the operating 

voltage. The variation in these parameters was observed while varying the cable power rating, the 

operating voltage, the configuration (AC or 50 percent redundant bipolar DC), and the AC frequency 

where applicable. The values sampled in this study are given in Table 19. 

The remaining assumptions applied in executing this sensitivity study are listed in Table 20. 

Cryogenic assumptions are given in Section 5.8. 

Several independent variables were selected to minimize the predicted mass of the overall system 

(including the cryogenic system). The optimized parameters are given in Table 21. Pipe flow diameters 

and critical current ratio both affect the cable weight via conductor and coolant mass. However, they also 

affect the cooling system requirements by changing the pumping power required and the cable losses. 
NASA/CR—2015-218440 88

Figure 79.—Dielectric Performance (Ref. 138). 

TABLE 19.—CABLE SENSITIVITY STUDY VARIABLE RANGES

Variable Values

Power, MW........................................................ 2, 4, 6, 8, 10, 12, 14, 16, 18, 20

Voltage, kV (RMS if AC, ±X if DC)................. 2, 4, 6, 8, 10, 15, 20, 25, 30, 35

AC or DC................................................................................................AC, DC

AC frequency, Hz ...................................................NA, 100, 200, 300, 400, 500

TABLE 20.—CABLE VOLTAGE SENSITIVITY STUDY ASSUMPTIONS

Variable Value

Superconducting material ................................................................ YBCO 

Conductor density..................................................................... 6380 kg/m3

Critical current density (Jc) ................................................ 2.00×1010 A/m2

Cooling medium ...................................................................................LN2

Temperature.........................................................................................70 K 

Final temperature .................................................................................77 K 

Coolant density ........................................................................... 810 kg/m3

Dynamic viscosity....................................................................... 150 μPa*s

Specific heat........................................................................... 2.14 kJ/kg*K

Stainless wall thickness...................................................................... 1 mm

Ambient temperature .........................................................................300 K 

Applied field (perpendicular)................................................................. 0 T 

Inner cooling flow diameter........................................................... 0.005 m 

Pipe roughness......................................................................................0.02

Safety factor (engineering Je/Jc) ............................................................. 0.1

Insulator dielectric strength (LPP)............................................. 50 kV/mm

LPP density................................................................................. 900 kg/m3

TABLE 21.—CABLE OPTIMIZATION VARIABLES 

Variable Range

Cooling flow pipe diameter, m.......................................(0,∞)

Critical current ratio........................................................(0, 1]
NASA/CR—2015-218440 89

In general, the mass and diameter sensitivities take the form of a unimodal curve with a minimum 

mass between 4 and 10 kV. The downward slope at very low voltages is generated due to increased 

conductor mass and thickness. The upwards slope at higher voltages is the result of increased insulation 

mass and thickness. 

5.3.11 DC Cable

The sensitivity results for the DC cable are given in Figure 80 and Figure 81. The geometry and mass 

of these cables were determined for a 30-m length cable. Overall cable and cryogenic system mass 

approximations are balanced taking the associated heat leakage from the environment into account. 

Figure 80 shows the mass and diameter sensitivity to voltage for the 12 MW DC case. This case is of 

interest due to power level. The power transmission requirement for the architecture of interest is 12.5 MW. 

In this case, the minimum cable weight per meter length is approximately 0.595 kg/m. This is achieved at 

around ±6 kV DC. 

Figure 81 shows the variation in optimal voltage as a function of overall power. The optimal voltage 

for a 2 MW cable is roughly 4 kV, while the optimal voltage for the 20 MW cable is approximately 10 kV 

shows the variation in overall cable diameter as a function of voltage for multiple power levels. 

Figure 80.—Mass and Diameter Voltage Sensitivities for a 12 MW 

capable LN2 Cooled Triax YBCO Bipolar DC Cable.

Figure 81.—Per Unit Mass Voltage Sensitivity for LN2 Cooled 

Triax YBCO Superconducting Bipolar DC Cable.
NASA/CR—2015-218440 90

Figure 82.—Diameter Sensitivity for LN2 Cooled Triax YBCO 

Superconducting Bipolar DC Cable.

Figure 83.—Per Unit Mass and Diameter Voltage Sensitivities for a 

12 MW capable LN2 Cooled Triax YBCO Superconducting ThreePhase AC Cable at a Frequency of 400 Hz.

5.3.12 AC Cable

The mass and diameter sensitivity results for the AC cables are given in Figure 83, Figure 84, and

Figure 85. The AC cable geometry and mass are determined assuming 1 m cable runs with their 

associated heat leakage from the environment. This significantly reduces the contribution of cryogenic 

systems during mass minimization. Considering the cooling system mass in the cable construction, the 

shorter the cable runs leads to a lower per unit mass.

As illustrated in Figure 83, the general trends for the 12 MW three-phase AC cable are similar the 

12 MW bipolar DC cables discussed previously. However, the optimal voltage shifts to ±15 kVrms. The 

optimal cable mass is also similar for the AC and DC cables. The mass per meter of the AC cable is 

approximately 0.576 kg/m. This reduction in per unit mass compared to the DC result is due to the ability 

to optimize cable design with little effect on cryocooling requirements (short length). 

Comparable to the bipolar DC case optimal voltage for the AC cable also shifts depending on the 

power level. This is illustrated in Figure 84. However, as the power level increases the optimal trend tends 

to become shallower; the system mass is less sensitive to the selected voltage between 8 and 25 kVrms. 

Figure 85 illustrates the sensitivity of diameter to voltage for the cable. This diameter is slightly larger 

in the AC case compared to the DC case due to the increased insulation thickness.
NASA/CR—2015-218440 91

5.3.13 Cable Losses

Heat leakage and conduction losses for superconducting cable systems are illustrated in Figure 86. In 

general, reductions in operating currents reduce the conduction losses for both AC and DC cable 

configurations. However, the heat leakage from the environment increase with voltage due to overall 

cable diameter increases. 

Figure 84.—Per Unit Mass Voltage Sensitivity for LN2 Cooled Triax 

YBCO Superconducting Three-Phase AC Cable.

Figure 85.—Diameter Sensitivity for LN2 Cooled Triax YBCO 

Superconducting Three-Phase AC Cable.

Figure 86.—Per Length Superconducting Conduction Losses and 

Environmental Heating as a Function of Voltage.
NASA/CR—2015-218440 92

Figure 87.—DC Cable and Support Systems Mass Contributions.

Figure 88.—DC Cable and Support Systems Mass Contributions.

The AC cable losses are over an order of magnitude higher than the losses for a DC cable. However, 

they are of the same general magnitude as the environmental leakage values. 

The heat leakage and conduction losses lead to increase in overall system mass from cryogenic 

cooling system. These contributions are illustrated in Figure 87 and Figure 88 for DC and AC systems, 

respectively.

Cooling systems are minor contributors to the overall systems mass. However, the results illustrated

here include optimal balancing of the critical current ratio and coolant flow cross sectional areas. 

5.4 Superconducting Magnetic Energy Storage

For the purposes of this sensitivity study, a superconducting magnetic energy storage (SMES) device 

was selected as the energy storage element. The advantages for this type of energy storage were summed 

up by Nielsen as follows (Ref. 142): 

• Capability of absorbing and delivering large amounts of power 

• High efficiency

• Long lifetime

• Short response time

• Completely static construction, low maintenance 

• All electric energy storage
NASA/CR—2015-218440 93

Within the TeDP architecture, the SMES will support bus stability and provide fill in power during loss of 

generating capability. This sensitivity model assumes air cored SMES in a force balanced coil (FBC) 

winding configuration (Refs..

143 and 144). An FBC SMES applies a winding method which allows for 

the management of forces within the coil.

The FBC winding configuration is shown in Figure 89. 

The size of the toroidal coil is defined in terms of its major and minor radii. These radii are shown in 

red in Figure 90. 

The ratio of major to minor radii affects the winding geometry and, in turn, the tensile and 

compressive forces acting on the SMES windings and structure (Ref. 145). 

5.4.1 Parameter Diagram 

Component sensitivity model I/O for the SMES is illustrated in the parameter diagram in Figure 91. 

Figure 89.—Force Balanced Coil Winding Configuration (Ref. 142). 

Figure 90.—Major (R) and Minor (r) Radii for the SMES Torus. 
NASA/CR—2015-218440 94

Figure 91.—Parameter Diagram for Energy Storage Device with Converter. 

5.4.2 Sizing Equations

5.4.2.1 Toroidal SMES Sizing

There are a number of article extending the analysis of Moon (Ref. 146) or quoting Lorentz force, 

current density crossed with magnetic field (Refs. 147 and 148).Tsutsui et al. applied the point version of 

Maxwell’s equations to an “object,” in the end being a toroid SMES. 

0

0

0

=⋅∇

μ=×∇

=⋅∇+×

B

jB

SBj

The Lorentz forces are not quite as straight forward as assumed by Tsutsui et al. due to the exclusion 

of magnetic field in the bulk of superconducting material. Only in relatively large magnetic fields, does 

either type 1 or type 2 superconductors allow penetration of their surface by magnetic field. 

Alternatively, ignoring the power source portion of the Poynting vector and integrating over a volume 

and time results in the energy within that volume for a magnetic material. Therefore, the fundamental 

expression which sizes the SMES device relates energy to inductance and current:

2

2

1 E = Li

where

E Energy stored in the SMES

L SMES inductance

i SMES current
NASA/CR—2015-218440 95

The inductance for a toroidal inductor can be calculated if the geometry and winding number are 

known: 





















 





+

+

+

μ=

2 2

2

22

1

22 rR

r

rR

r NL

where

N number of turns 

μ SMES inductance

i SMES current

R major radius of the torus

r minor radius of the torus 

Substitution of the inductance equation for a toroidal coil into the energy equation for the SMES 

yields:

( ) ( ) ( )2

2 2

2

1

4

1 Ni

rR

r

rR

r E 































+

+

+

μ =

This equation can be expressed as a function of the ratio of torus radii ( ) r

R =α instead of the minor 

radius: 

( ) ( ) ( )2

2

12

1 1

14

Ni R E 































+α

+

+αα

μ =

The product of N and i in this expression can also be expressed in terms of the SMES magnetic field 

strength: 

μ

π

=⇒

π

μ=

BR Ni

R

Ni B 2

2

This yields an expression for energy in terms of the major radius, the magnetic field density, and the 

ratio of torus dimensions: 

( ) ( ) 































+α

+

+ααμ

π =

2 22

3

12

1 1

1

B 1 RE

Thus, the size of the magnetic coil can be expressed as a function of the energy storage required, the ratio 

or torus radii, and the magnetic field strength. 
NASA/CR—2015-218440 96

( )

( ) 

































 







+α

+

+αα 



 





π

μ = 22 2

3

12

1 1

1

B

ER

For typical values for α (ranging from 4 to 6 (Ref. 149)), an approximation for major radius can be used. 

( ) 1 22

3  +αα

 





π

μ ≈ B

ER

5.4.2.2 Superconductor Mass Equations

The mass of the SMES superconducting material is a function of the length, density, and crosssectional area of the superconducting windings. The cross-sectional area is a function of current and 

critical current density.

e

scscscsc scscsc J

i

ρ= ρ=⇒ lmalm

The number or windings is calculated from the magnetic field density equation and a simple relationship 

between the SMES power required and the operating voltage (P = iV). 

P

VBR N

μ

π = 2

The length of the superconducting windings can then be calculated considering the toroidal geometry. 

2

2 2 



 



 +π=

N

R rNlsc

Combining these equations gives an expression for superconductor mass.

2

1

2

2

2

2

4





















 





π

μ +

μ

π

ρ= V

P

B

r

J

BR

m e

scsc

From this expression we see that the mass of the windings is a weak function of operating voltage. 

However, for typical SMES devices the R/N value is negligibly small. Therefore, the expression can be 

simplified to 

e

scsc J

BR

m

μα

π

ρ≈

2 4

5.4.2.3 Structural Weight Calculation 

Magnetic energy storage in a toroid inductor results in compression for the minor radius and tension in 

the major radius. This seems to contradict with some other research. Yet, even that research seems to 
NASA/CR—2015-218440 97

diverge from later experimental research (Refs. 150, 151, 152, 153, and 154). Upon further review of 

literature, the fundamental Lorentz forces for superconductors seem to be in international discussion, with a 

conclusion not in sight. Of which, none conveniently match the normal conductor Lorentz force description, 

i.e., the electrical version of Newtonian versus Einsteinian. One researcher’s experiments indicate that the 

relative mechanical force measured approaches zero when conventional definition indicates maxima 

(Ref. 155). Due to the ongoing research defining superconductor electromagnetic forces.

Moon’s estimation was chosen for our estimations. We assumed that the “quality factor,” which is a 

description of compression to tension relationship, is optimal. Assuming tension, a Zylon epoxy type 

material ratings are used. Virial theorem based SMES structural mass estimates do not show sensitivity to 

operating voltage (Ref. 156). The main drivers are the energy stored in the device and the material 

properties of the structure. 

( )

str

str

str

E

m Q

σ

ρ += 21

The Q value in this expression is a ratio of tensile to compressive strength (Ref. 157). It is expected that 

this value will be sensitive to the ratio of torus radii. However, without a clear relationship between 

SMES geometry and Q value, the sensitivity results sample Q on its range from 0 to 1. 

5.4.2.4 Dielectric Insulation Weight Calculation

It is assumed that the dielectric insulation is a function of the surface area of the torus (ATorus), the 

operating voltage (V), and the number of insulation layers required (n).

( ) n

V Am

ins

Torusinsins

κ

ρ=

The equation for the surface area of a torus is given by:

ATorus Rr 2 4π=

The number of insulation layers depends on the number of winding layers are needed. The inner 

circumference of the torus, number of windings, and the geometry of the superconductor are all factored 

into determining how many layers of windings are required. This number is calculated with this equation: 

( )



 





−π = rR

Nw roundupn sc

2

5.4.3 Power Electronics

SMES charge and discharge operations are facilitated by an H-bridge circuit (Ref. 158). This is 

illustrated in Figure 92 with the operating mode power flows: charge, steady state, and discharge. Power 

flow is maintained with two diodes and two modulated IGBTs. The control of these IGBTs determines 

the operating mode for the SMES system.

Diode, IGBT, and capacitor parametric sizing and efficiency models use the converter device trends 

discussed in Section 5.1. Output capacitance requirements require assumptions regarding the duration 

over which the capacitor can maintain voltage.
NASA/CR—2015-218440 98

Figure 92.—SMES H-Bridge with Mode Power Flow Superimposed.

The only device illustrated in Figure 91 not modelled in the sensitivity study is the dissipative 

element placed in parallel to the SMES inductor. In the case of a SMES loop fault condition, this arresting 

device may be required to overcurrent while the energy can be dissipated during fault conditions. High 

power dense power dissipation device alternatives and fault tolerant SMES configurations need to be 

explored further to enable implementation in the TeDP architecture.

5.4.4 Cryocooling Weight Calculations

A recirculated layer of cryocoolant is required to maintain superconducting temperatures. The SMES 

toroidal inductor introduces some heat from conduction losses. Additionally, some heat is transferred to 

the coolant from the environment through the insulation. A conformal tank is used around the SMES 

toroid with flow of coolant around the SMES toroid as illustrated in Figure 93. 

The coolant channel is sized to minimize the mass of the overall system by considering the pumping 

requirements and the mass of the coolant. The coolant layer is contained in a vacuum jacketed insulation 

system with an assumed structure as introduced in Section 5.3.5. 

5.4.5 SMES Sensitivity Study Parameters and Assumption Overview

SMES sensitivity study variables ranges are provided in Table 22, and SMES voltage sensitivity 

study assumptions are provided in Table 23. 

The overall system mass is optimized by varying the independent parameters given in Table 24. 

Cryogenic assumptions are given in Section 5.8. 

TABLE 22.—SMES SENSITIVITY STUDY VARIABLE RANGES

Variable Values

Aspect ratio.....................................................................3, 4, 5, 6

Magnetic field................................................................5, 7.5, 10

Energy stored, MJ....................................... 30, 60, 100, 150, 200

Operating voltage, kV....................... 2, 4, 6, 8, 10, 12, 15, 20, 25

Compressive quality factor ............................ 0, 0.25, 0.5, 0.75, 1

Cryocoolant...........................................................LN2, LHe, LN2
NASA/CR—2015-218440 99

TABLE 23.—SMES VOLTAGE SENSITIVITY STUDY ASSUMPTIONS

Variable Value

Superconducting material .................................................. YBCO 

Conductor density, kg/m3......................................................6380

Insulator dielectric strength (LPP), kV/mm..............................50

LPP density, kg/m3 ................................................................. 900

Structural material.......................................................Zylon fiber

Structural material density, kg/m3.........................................1560

Output capacitor voltage support duration, s .............................. 1 

Ultimate tensile strength, GPa .................................................5.8

TABLE 24.—SMES OPTIMIZATION VARIABLES

Variable Range

Inlet coolant temperature, K (Depends on medium)

Critical current ratio (0,1)

Number of Mylar insulation layers [0, ∞)

Coolant delta T, K (Depends on medium and inlet temperature)

Coolant layer thickness, mm [1, ∞)

Figure 93.—SMES Coolant Flow.

5.4.6 Mass and Efficiency Trends

The mass of the structural and superconductive elements are insensitive to voltage. However, increase 

in voltage does affect the insulation weight due to increased off-stand voltage required and the numbers of 

layers of insulation. As such, assuming a fixed power requirement, the maximum energy density of the 

SMES coil is achieved on a SMES device by minimizing the operating voltage. To provide a fixed power, 

a reduction in voltage requires an increase current. Therefore, the induction requirement is reduced and 

the number of windings decreases. 

The low voltage power density of the SMES coil must be balanced with power density of the power 

electronics and cooling systems both of whose power density increases with voltage. The sensitivity of 

the SMES to energy and voltage are illustrated in Figure 94. All results in this section apply a power 

requirement of 12.5 MW supplied by the SMES. 

Figure 95 and Figure 96 illustrate the overall SMES system mass and energy density as a function of 

the amount of stored energy and operating voltage. 
NASA/CR—2015-218440 100

Figure 94.—Sensitivity of Overall SMES System Mass to Voltage.

Figure 95.—Sensitivity of SMES Mass to Voltage and Energy Considering 

Power Electronics and Cooling Systems.

Figure 96.—Sensitivity of SMES Energy Density to Voltage and Stored 

Energy Considering Power Electronics and Cooling Systems.
NASA/CR—2015-218440 101

Figure 97.—Sensitivity of SMES Energy Density to Voltage and Stored Energy for Structural 

and Electric Components Only.

The primary driving trends for voltage selection for this system are the power electronics and 

cryocooling systems. The power density of the SMES toroid without the support systems is illustrated in 

Figure 97. 

5.4.6.1 Toroidal Inductor Mass Sensitivities

Operating voltage has a large effect on the number of windings on the SMES. The mass of the 

superconducting material may by insensitivity to this voltage (Figure 98). However, the winding 

complexity increases dramatically with increased voltage. An increased number of windings requires 

additional layers of insulation required for dielectric protection (Figure 99).

Field, aspect ratio and the compressive quality factor play a lesser role in the SMES sensitivity. These 

trends are illustrated in Figure 100, Figure 101, and Figure 102. 

The energy density of the SMES inductor is also sensitive to the operating temperature. By assuming 

the use of YBCO as the superconducting material for SMES windings, the superconducting temperature 

range is quite large. However, the critical current density is affected greatly by the operating temperature. 

This effect is illustrated in Figure 103. 

The primary loss mechanisms during SMES discharge are driven by conversion and power 

electronics cooling. During standby operation, a fully charged SMES toroid exhibits losses as illustrated 

in Figure 104. These losses are driven by conduction losses in the superconductor and heat leakage from 

the environment. 

During SMES charge and discharge at 12.5 MW, the power loss is much higher as illustrated in 

Figure 105. 
NASA/CR—2015-218440 102

Figure 98.—Sensitivity of the Number of SMES Windings to Voltage and Stored Energy. 

Figure 99.—Number of Layers of Superconductor Windings.

Figure 100.—Mass Sensitivity of SMES Structural and Electrical Elements to Voltage and Magnetic 

Field Density.
NASA/CR—2015-218440 103

Figure 101.—Sensitivity of SMES Energy Density to Voltage and Torus Aspect Ratio.

Figure 102.—Sensitivity of SMES Energy Density to Voltage and Compressive Quality Factor.
NASA/CR—2015-218440 104

Figure 103.—Sensitivity of SMES Energy Density to Voltage and Temperature.

Figure 104.—Standby Losses for Fully Charged SMES Toroid.
NASA/CR—2015-218440 105

Figure 105.—SMES System Losses during 12.5 MW Charge/Discharge.

5.5 Circuit Breakers

The current interruption devices serve as the fault isolation devices for this architecture. Sections 3.4 

and 4.0 discuss the types and trade-offs of electromechanical, hybrid, and solid-state circuit breakers.

Schematics of examples of these types of circuit breakers are illustrated in Figure 106. Based on prior 

studies and simulations of DC microgrid systems (Refs. 159, 160, and 161), the time from the fault 

initiation to the fault current peak magnitude for this superconducting system will likely occur in several 

microseconds. The resulting fault energy will be subject to the system component’s quench transition. If it 

takes longer for the protection equipment to interrupt the fault current, the equipment will have to be 

oversized to have a higher fault current rating, the circuit breakers will have to have higher interruption 

ratings and a larger cryocooler mass will be need to dissipate the fault energy generated during an event.

If the circuit breaker can interrupt the fault current before the peak fault current magnitude, then the rest 

of the faulted system will only have to withstand a reduced peak current magnitude. The desire to 

minimize size and weight of the electrical components and cryocooler necessitated a very fast interruption 

time and based on the interruption times of the time of the considered circuit breakers, the solid-state 

circuit breaker was chosen for this study. This was based on the assumption that the protection system 

should be designed interrupt the fault current as early as possible and limit the fault current, to twice 

nominal current in this study. The schematic of the modeled SSCB is shown in Figure 106C. Other hybrid 

and SSCB topologies were previously discussed in Sections 4.3 and 4.4. 

Additional interrupter models of an electromechanical and/or hybrid circuit breaker could be 

developed and used with system trade studies to determine the impact of the current interruption device 

on the system weight and efficiency. 

The following sections describe the DC SSCB. An AC SSCB model would use similar scaling of the 

SSCB components, but the ratings of these components would differ, and the losses of the SSCB would 

differ.
NASA/CR—2015-218440 106

Figure 106.—Current Breaker Device Topologies. Solid-State Circuit Breaker Selected for Modeling.

5.5.1 Model Overview

5.5.1.1 Parameter Diagram 

The parameter diagram shown in Figure 107 outlines the SSCB model major inputs, outputs, and 

control parameters. For a given rated power based on the SSCBs placement in the architecture and the 

desired DC voltage for study, the model estimates the SSCB mass and losses. Other control parameters 

used to scale the SSCB are the operating temperature and topology. For this study, the topology is fixed 

for the topology shown in Figure 106C. Intermediate outputs of the SSCB model are its current 

interruption time and fault current rating based on IGBT scaling. These outputs will be used by the 

protection system to coordinate SSCB and SFCL interaction. Instead of these values computed as an 

output of the SSCB model, they could be specified as inputs to the SSCB model by the protection system. 

The SSCB model is structured similarly to the inverters and rectifiers. Given the specified rated 

power and DC voltage, the component voltage and current ratings are specified. From these component 

specifications, the component masses and losses are computed and totaled. As indicated by Figure 106C, 

the SSCB components are the conduction diodes, IGBTs with freewheeling diodes, and a metal oxide 

varistor (MOV). The MOV is used as an overvoltage protection device during switching events. 

5.5.1.2 Governing Equations

Several equations are used to determine the component ratings. The nominal DC current rating is 

calculated by

ptpdc

dc

dc V

P I ,

=

Given these ratings, the IGBT component voltage and current ratings are calculated by: 

s

ptpdc

block IGBT N

V

V 1

2

, , 













 =

p

dc

IGBT N

I I =

The IGBTs are arranged in series and/or parallel in a similar manner as for the IGBT power converter 

scaling in order to maintain interpolation of the IGBT scaling data. The IGBT scaling data is the same for 

both the power converter and SSCB IGBTs. 

CB: Mechanical

CB: Hybrid CB: Solid-State

Load-breaking switch

A B C
NASA/CR—2015-218440 107

Figure 107.—Parameter Diagram for Current Interrupter. 

The voltage rating is determined by half of the DC bus pole-to-pole voltage since a SSCB is used on 

both of the DC poles and will be required to block the single pole-to-ground voltage. 

These voltage and current ratings are used by the IGBT, diode, and varistor component sizing model 

to estimate the mass and losses of these devices. The IGBTs are scaled based on room temperature and 

cryogenic test data in the same way as for the power converters. For this topology, two diodes also 

normally conduct along with the IGBT. The cryogenic diode scaling is the same as previously described 

in Section 5.2.2.2.2. The conduction diode mass is estimated as a percentage of the IGBT mass based on 

the ratio of an individual diode mass to a similarly rated IGBT and freewheeling diode mass. The model 

could be expanded to create an independent conduction diode scaling model and mass estimate.

Only the conduction losses are considered to estimate the normal operational efficiency of the SSCBs. 

During a switching event, the switching losses could also be estimated, but these losses are not part of the 

normal SSCB operation. Any type of DC circuit breaker will incur switching losses to interrupt current, 

but different types of DC circuit breakers have different conduction losses. The purpose of this model is 

to estimate the SSCB conduction losses during normal operation. The IGBT and diode conduction losses 

are estimated by equation 

( )DIrIVPcond CE C CE C

2 +=

where the duty cycle is equal to 1. 

The total IGBT and diode losses are calculated by 

( ) loss = PP cond,IGBT + 2P , NN psdiodecond

Note that the SSCB can operate bidirectionally. This capability results in a SSCB with almost twice the 

number of components as a unidirectional SSCB. Only half of the components conduct during normal 

operation. 

The MOV mass is estimated from existing surge arrester masses and the ratio of those masses to 

IGBT masses with similar voltage rating. Surge arrester data from ABB for devices rated for 1 to 2 kV 

and 2.5 to 4.7 kV were gathered (Refs. 162 and 163). The MOV mass estimate is calculated by:

SSCB

Weight

Heat Quality

Efficiency

Current Interruption Time

Type and Topology

DC Voltage

Rated Power

Temperature

Fault Current Rating
NASA/CR—2015-218440 108















>

≤

=

, kV2

, kV2

, ,

,2

,2

, ,

,1

,1

,

IGBT scaled block IGBT

IGBT single

MOV single

IGBT scaled block IGBT

IGBT single

MOV single

MOV stotal

M V

M

M

M V

M

M

M N

Where MMOV1,single is the surge arrester mass data for a voltage range of 1 to 2 kV operation, MIGBT1,single is 

the average IGBT mass for the voltage rating 1 to 2 kV, MMOV1,single is the surge arrester mass data for a 

voltage range of 2.5 to 4.7 kV, MIGBT2,single is the average IGBT mass for the voltage range 2.5 to 4.7 kV, 

and MIGBT,scaled is the IGBT mass estimate per the model IGBT scaling. 

Further understanding of varistor operation and material properties at cryogenic temperatures is 

needed to adequately scale the varistor at different operating temperatures. It is assumed that no 

significant losses are incurred by the varistor during normal operation. 

The current interruption time is estimated from the IGBT scaling for the specified voltage and current 

ratings as described by Section 5.2.2.2.2 and Figure 42. 

5.5.2 Mass and Efficiency Trends

5.5.2.1 High Power SSCB Trends

The high power DC SSCBs are rated at 12.5 MW which is 50 percent of the minimum takeoff power 

25 MW. The following trends in Figure 108, Figure 109, and Figure 110 were developed from the 

equations and assumptions discussed in Section 5.5.1.2 and tabulated in Table 25. 

TABLE 25.—ASSUMED PARAMETERS FOR SSCB MODEL TRENDS 

Parameter Value

Converter Sizing Operating Temperature....................................................... 100 K

Conduction Diode Mass Scaling Ratio Diode/IGBT............................................. 0.9

Varistor Mass Scaling Ratio MOV/IGBT, ≤ 2 kV ........................ 0.76923

Mass Scaling Ratio MOV/IGBT, >2 kV ......................... 0.65011

Figure 108.—High Power SSCB Current Interruption Time versus DC Bus Voltage. 
NASA/CR—2015-218440 109

Figure 109.—High Power SSCB Quantity of Series and Parallel Switches versus DC Bus Voltage.

Figure 110.—High Power SSCB Efficiency versus DC Bus Voltage.

Figure 111, Figure 112, and Figure 113 show the high power SSCB mass trends as a function of DC 

bus voltage. The minimum mass of 33.9 kg (maximum specific power of 329 kW/kg) for this high power 

SSCB is found at a DC bus voltage of 6.5 kV. Note that the discontinuities in these trends are due to the 

IGBT configuration in parallel and in series in order to interpolate the IGBT data for scaling. Figure 109 
NASA/CR—2015-218440 110

shows the number of IGBTs in parallel and series used to form each solid-state switch for each DC bus 

voltage. 

The trends in mass versus DC bus voltage can be further interpreted by analyzing the mass 

breakdown versus DC bus voltage as shown in Figure 114. The diode mass contributes to a slightly larger 

percentage of the total SSCB mass for lower voltages, while the varistor mass contributes a higher 

percentage of the total mass for higher voltages. However, the mass trends are dominated by the IGBT 

mass sizing. 

Figure 113 and Figure 114 show the high power SSCB efficiency and conduction losses. These trends 

show an improvement in efficiency for high DC bus voltages. The losses continue to decrease for 

increasing voltages up to 6.5 kV to achieve an efficiency of 99.77 percent at 6.5 kV. For voltages greater 

than 6.5 kV, the efficiency and losses remain approximately the same. The conduction losses are 

determined in part by the IGBT conduction voltage drop scaling as a function of the rated IGBT voltage. 

Figure 111.—High Power SSCB Conduction Losses versus DC Bus Voltage.

Figure 112.—High Power SSCB Specific Power versus DC Bus Voltage.
NASA/CR—2015-218440 111

Figure 113.—High Power SSCB Mass versus DC Bus Voltage.

Figure 114.—High Power SSCB Mass Breakdown by Component versus DC Bus Voltage.

Figure 114 shows the high power SSCB current interruption time trends versus DC bus voltage. This 

trend is similar to the power converter interruption time trends due to similar scaling of IGBT turn-off 

times. The discontinuities for lower voltages are due to the parallel switch configurations. Generally for 

higher voltages, the interruption time decreases. 
NASA/CR—2015-218440 112

5.5.2.2 Low Power SSCB Trends

The low power SSCB trends that follow in Figure 115 to Figure 121 were developed from the SSCB 

at a rated power of 2.5 MW which is 7.14 percent of the minimum take-off power. These trends are 

developed with the same assumptions as listed in Table 25. Note that according to the selected 

architecture, the low power SSCBs are rated at either 5.36 percent (1.88 MW) or 7.14 percent of the 

minimum takeoff power. The trends for the 1.88 MW SSCBs are not included in this report but are 

similar to the 2.5 MW SSCB trends. 

Figure 115.—Low Power SSCB Mass versus DC Bus Voltage.

Figure 116.—Low Power SSCB Mass versus DC Bus Voltage. 
NASA/CR—2015-218440 113

Figure 117.—Low Power SSCB Conduction Losses versus DC Bus Voltage. 

Figure 118.—Low Power SSCB Efficiency versus DC Bus Voltage. 
NASA/CR—2015-218440 114

Figure 119.—Low Power SSCB Series and Parallel Configuration versus DC Bus Voltage. 

Figure 120.—Low Power SSCB Current Interruption Time versus DC Bus Voltage. 
NASA/CR—2015-218440 115

Figure 121.—Low power SSCB Mass Breakdown by Component versus DC Bus Voltage. 

Figure 115 and Figure 116 show the low power SSCB mass trends versus DC bus voltage. The 

minimum mass for this trend is 10.70 kg at 5 kV, yielding a maximum specific power of 149.4 kW/kg. 

Note that the discontinuities follow the series and parallel configuration of the IGBTs as shown in 

Figure 119. Also, the data developed for these trends beyond 7.5 kV required extrapolation of the IGBT 

scaling data. Because the power rating is lower, the minimum mass for the 2.5 MW SSCB is lower than 

for the 12.5 MW SSCB, but the specific power of the lower power SSCB is about half that of the higher 

power SSCB. Looking at the SSCB component mass breakdown in Figure 121, the mass of each 

component for the lower power SSCB has a similar percentage of the total SSCB mass as for the higher 

power SSCB. So, the specific power difference is not attributed to the component scaling. The low power 

SSCB specific power difference is attributed to the IGBT mass scaling for voltage and current. For 

similar voltages but much lower currents, the IGBTs cannot process as much power per kg as IGBTs 

rated for higher currents. This increases the high power SSCB specific power. 

Figure 117 and Figure 118 show the conduction loss and efficiency trends for the low power SSCB. 

The efficiency increases to 99.75 percent for increasing voltage up to 6 kV and then maintains a relatively 

constant efficiency. This efficiency is similar to the high power SSCB. 

Figure 120 shows the low power SSCB current interruption time as a function of bus voltage. The 

current interruption time generally decreases for higher voltage. The rate of range of the decrease in 

interruption time is greater from 1 to 15 kV than for voltages greater than 15 kV. Note that the 

interruption times greater than 7.5 kV are the result of IGBT scaling data extrapolation. 

The trend profiles for both the high and low power SSCBS are similar. A notable difference is that the 

specific power of the low power SSCB is approximately half that of the high power SSCB. The high 

power SSCB power rating is 7 times higher than the low power SSCB. The efficiency of the high and low 

power SSCBs are similar. The DC bus voltages for the high and low power SSCB minimum weight and 

maximum efficiency are similar.
NASA/CR—2015-218440 116

5.6 Superconducting Fault Current Limiters

The role of superconducting fault current limiters (SFCL) in this TeDP system is, first, to provide 

quench protection for the overall system, and secondly to reduce the current requirements for interruption 

equipment. 

With regard the first protection role; the fault current limiter is designed to be the first and only piece 

of the system to respond to an overcurrent via quench. Other equipment will be exposed to an elevated 

during the faulted conditions. However, they will not be required to manage internal quenches. 

The second protection role of the fault current limiter is to reduce the magnitude and duration of the 

overcurrent experienced by the remaining equipment. This leads to a reduction in equipment weight. 

5.6.1 Modeling Approach

There are many different types of superconducting fault current limiters. For the purposes of this 

study the resistive type SFCL was selected.

The resistive type fault current limiter has some distinct advantages. Chiefly, by relying on the natural 

resistivity transition properties of superconductors, it has the simplest design and construction. 

Additionally, the resistive SFCL tends to be the lightest solution. For these reason this option is the most 

widely implemented method for conventional terrestrial systems.

This transition between superconducting and non-superconducting states is illustrated in Figure 122. 

This figure shows a generic log-log relationship between the electric field generated by a superconductor 

and current density. Three states are indicated: flux-creep state, flux-flow, and resistive. As the current 

increases in a superconductor the losses increase. Once the transition to flux-flow occurs, thermal effects 

in the superconducting material further increase the generated electric field losses.

The key to resistive SFCL protection design is balancing the current capability of all system 

components with the quench current (Figure 123). Using linear approximations between temperature and 

resistivity during superconducting transition, Blair et al. analytically solve for quench transition time for 

BSCCO round wire in a DC system (Ref. 164). His results are illustrated in Figure 123. 

A similar relationship between fault current and quench time can be envisioned for each superconducting component within the system. Proper protection is provided by ensuring the quench time/fault 

current curves for all equipment sit substantially to the right of the curve to the SFCL. 

Figure 122.—Generic E-J Characteristic Superconductor (Ref. 165). 
NASA/CR—2015-218440 117

Figure 123.—SFCL DC Quench Time Analytical Solution from Reference 164. 

5.6.2 Geometry and Winding

The reference configuration for this fault current limiter voltage sensitivity study is a prototype unit 

developed in Manchester University (Refs. 166, 167, and 168). This SFCL unit consists of helical wound 

MgB2 wire around a slotted alumina former. This unit is shown in Figure 124 and Figure 125. 

The Manchester unit applied two interleaved solenoid coils which were wound with opposing 

currents to cancel out solenoid inductive fields. Additionally, as seen in the two figures, copper braid was 

also situated on the former to allow for transitioning between superconducting and normally conducting 

wires. This entire former, winding assembly is placed in a cryostat bath.

For the purposes of this component sensitivity exercise, Manchester’s general solenoidal construction 

method is applied. However, the Manchester University resistive type SFCL coil was wound and operated 

in a manner which reduced inductance generated by the superconducting coil. In contrast, for this DC 

microgrid, operating the fault current limiter as an inductor is beneficial to the protection system 

performance. Therefore, the desired inductance provided by the SFCL coil determines impacts geometry 

of the solenoidal coil. 

The spacing between each winding is determined to achieve the desired inductance, and minimize 

weight, while ensure adequate dielectric protection and limit the magnetic field in the core. Additionally, 

alumina is replaced with a glass reinforced plastic (GRP) composite to further minimize the former 

weight. 

The general arrangement for the notional SFCL developed for these sensitivity models is illustrated in 

Figure 126. 
NASA/CR—2015-218440 118

Figure 124.—University of Manchester SFCL Windings on Slotted Former (Ref. 168). 

Figure 125.—University of Manchester SRCL Former Drawing (Ref. 168). 

Figure 126.—General Arrangement of a Notional Vacuum Cored Solenoidal SFCL 

used for First Principles Based Mass and Efficiency Sensitivity Modeling.

Insulated

Superconducting

Windings

Vacuum Cryostat

Former

Insulated Windings

Coolant Reservoir

Vacuum 

Core

Vacuum Core

Vacuum

Core
NASA/CR—2015-218440 119

5.6.3 Parameter Diagram

The SFCL model inputs and outputs follow the structure illustrated in Figure 127. The primary 

attributes of interest are the weight and efficiency of the fault current limiter. Additional outputs provide 

information to determine protection system effectiveness. SFCL superconducting, and quench state 

resistance, as well as the quench transition time are reported out to determine the expected overcurrent for 

the system. 

To size the SFCL, information about the nominal operating current and the quench to nominal current 

ratio are required. Additionally, the protection system must specify the inductance desired from this 

solenoidal coil and the time expected between fault occurrence and fault interruption. Finally, the 

operating voltage is specified to size the required dielectric insulation.

There are many internal variables that affect the sizing trends for this device. This includes the 

coolant temperatures for nominal operation, resistance transitions parameters required from Blair’s 

quench time models, material assumptions for former and dielectric, and the geometry and flux density 

limits for the solenoidal sizing. 

Figure 127.—Parameter Diagram for Superconducting Fault Current Limiter. 

SFCL Quench Transition Time

Superconducting State R

Quench to Nominal

 Current Ratio

Nominal Rated Current

SFCL 

Weight

SFCL Efficiency in 

Superconducting 

and Quench States

Desired L

Quench State R

Temperature

Time to Isolation

Voltage

L

Max Inductor B Field

Geometry Limits ...
NASA/CR—2015-218440 120

5.6.4 Governing Equations

The governing equations for the fault current limiter model are presented in four sections: quench 

transition modeling, inductor modeling, mass modeling, and solver overview. 

1. The quench transition modeling equations determine the time to quench for the SFCL device as well 

as the heat that is dissipated to the cryocoolant. 

2. The induction equations capture the relationships between the physical geometry of the cable coil and 

which determines the impedance properties of the SFCL.

3. The mass equations take the geometry properties and the dissipated heat information to determine the 

overall mass of the inductive coil and the coolant.

4. The solver routine determines the geometry parameters of the SFCL required to minimize the mass of 

the device.

5.6.5 Quench Modeling

The SFCL sensitivity model applied the processes developed by Blair et al. to determine the time to 

quench (Ref. 169). This process makes a fundamental assumption that the resistivity of the 

superconductor varies linear with temperature during a quench event when the temperature is greater than 

over the critical temperature.

This assumption is reflected in the induced field and critical current density equations. Blair presents 

these equations thusly: 

( ) ( )

( ) ( )















−

− ≈

















ρ≈

0

0

,

TT

tTT JTJ

tJ

T

T TtE

C

C

C c

C

where

E electric field

ρ normally conducting resistivity 

T superconductor temperature

J current density 

JC critical current density

TC critical temperature

T0 reference temperature

Jc0 JC at reference temperature

t time

The next equation used to determine quench time is an expression for electric field sensitivity to 

temperature developed by Paul et al. (Ref. 170):

( ) ( )

( ) β β















































 = 0

0

0

0

77

C c

c n c

J

tJ

TJ

J

E

E ETE k
NASA/CR—2015-218440 121

where

E0 field at flux flow to flux creep transition 

Ec E at critical current density

T temperature

β flux flow region exponent

n77K flux creep region exponent at 77 K reference temperature

The final expression needed to solve for quench time is the superconductor material temperature: 

( ) [ ( ) ( )] ∫ += − dttQtQ

c

TtT sc

cs

a coolant

1

where

Ta coolant reservoir initial ambient temperature 

csc superconductor heat capacity

Qsc heat dissipated in the superconductor 

Qcoolant heat removed by coolant reservoir

Combining the linear assumptions and the field transition expressions, and substituting into the 

thermal equation, the time to quench can be solved by the following equation: 

∫





 





θ

− − −

− =

sc

amb

c

c init

sc TT

TT

TTk

dT

ct

where k is defined by 

sc

C sc

n c l

aJ

I

E

E Ek 





























 = +β

β

0

1

0

0

0

and where 

lsc length of the superconducting wire

asc cross-sectional area of superconducting wire

θsc thermal resistance from superconductor to reservoir

Blair et al. provides an analytical solution the integral expression for quench time. 

5.6.6 Quench Cooling

The quench transition time equations assume that the overall quench energy is only partially 

dissipated to the coolant reservoir, while the remainder acts to heat the superconducting material. For this 

estimate, the heat transfer coefficient between the cable and the reservoir to determine the thermal energy 

dissipated to the coolant during the quench event. 

However, a conservative estimate for coolant reservoir sizing assumed that the entire quench energy 

is dissipated to a fixed volume of coolant. Under this assumption the total amount of heat assumed the 

following expression: 
NASA/CR—2015-218440 122

( ) ( ) ∫ = isolate

fault

2

coolant

t

t Q q dttRti

where

Qcoolant Heat removed by coolant reservoir

iq Quench current

R Resistance in superconducting wire

Applying linear superconducting to conduction resistivity transition assumption, the overall heat 

which must be dissipated is a function of the time to quench and the time it takes to interrupt the current. 

Therefore the overall heat dissipated in the cooling plenum is given by: 

( )















+ − ++= p q

q

sc sc q

q ttRt

t

R RR

iQ isolate isolate

2

coolant max ,0 22

where

Rsc superconducting state resistance

Rq quench state resistance

R resistance in superconducting wire 

tq time to quench 

tisolate time to isolation

This heat value is used to determine the mass and volume of the coolant reservoir. 

5.6.7 Induction Equations

Solenoid inductance equations were used to size the SFCL coil geometry. It should be noted that this 

set of equations is only valid for coils where the diameter is much smaller than the length. Additionally,

air core assumptions were applied for the field and inductance equations for these SFCL sensitivity 

models. Therefore the relative permeability was assumed to be roughly unity. 

Determining the performance for the optimal SFCL inductor design required that the mass be 

minimized in terms of inductor geometry which still providing the requirement functionality and concept 

feasibility. The optimization followed the formulation below:

( )

min max

max

total

, ..

min

LLL

db

d

l

a

BB

ts

m

Nd

F

F

F

F

≤≤

≤

≤

≤

where

dF solenoid diameter

N number of solenoid windings

B,Bmax magnetic flux density and limit 

lF solenoid length

a,b solenoid geometry limits

L,Lmin,Lmax solenoid inductance and limits
NASA/CR—2015-218440 123

Several limits were applied to in this solenoid inductance model. Firstly, the spacing between 

inductor windings was limited to ensure that the induced magnetic field was limited to a desired value. 

Variation of SFCL performance with respect to the limiting B field is captured in the component 

sensitivity portion of this section. Magnetic flux density for an air core solenoid inductor is given by: 

F

q

l

Ni

Bmax μ≥

where

Bmax maximum allowable magnetic flux density 

μ magnetic constant 

N number of solenoid windings 

iq quench current

lF length of solenoid

The length of the solenoid is directly proportional to the product of the number of windings and the 

winding spacing: 

lF = Nh

where

h solenoid winding spacing 

Substituting this relationship into the magnetic flux density relationship provides a constrained 

relationship between winding spacing and magnetic field. 

The next set of limits applies to the geometry of the solenoid former. Minimum geometric limits on 

solenoid diameter (dF) and length (lF/dF) are applied in this voltage sensitivity model which can be 

manipulated to ensure validity of the solenoid inductance equations. For more accurate estimation and 

sizing, a series of inductance equations could be implemented in future model iterations which are 

applicable to various inductor geometries (solenoid, pancake, toroidal, etc.) 

The final constraint on the SFCL geometry comes from the protection system requirements. With 

limits on the maximum allowable flux density and limits on the solenoid diameter and length, the 

inductance value can be overconstrained if the solenoid winding spacing is fixed. Therefore, the 

inductance value was allowed to float so long as it meets a set of minimum requirements and does not 

affect system performance by being too large. 

max

2 2

min 4

L

l

dN L

F

F ≤ π μ≤

where

Lmin minimum required inductance

Lmax maximum required inductance 

dF solenoid diameter

If the inductance value is not allowed to float above the minimum value, than the solenoid winding 

spacing value varies depending on the specified inductance.
NASA/CR—2015-218440 124

Winding spacing is therefore subject to three limits: cable geometry, magnetic field, and inductance. 

The winding spacing must be larger than the insulated cable diameter. It must also be large enough to 

ensure that the magnetic field does not exceed the requirement. Finally, it must be large enough to ensure 

that the desired inductance is equal to the value specified.















 π

μ+

κ = L

dN

B

i

d V h q F sc

ins 4

max ,,2

2

max

where

κins dielectric strength of the insulation

V cable voltage

dsc diameter of the BSCCO round wire 

The cable diameter in this equation assumes that the insulation is wound around the BSCCO round wire. 

The efficiency of the SFCL is a simple function of the length of the superconductor cable (lsc). This 

length is calculated in terms of the inductor geometry as a helical coil:

2 2 2 hdNlsc F +π=

5.6.8 Mass Equations

The overall system mass is a combination of the masses of the superconducting wire (msc), the 

solenoid former (mF), the dielectric insulation (mins), and the coolant (mcoolant). While the cryostat 

contributes greatly to the mass of the SFCL devices, this mass estimate is not currently included in these 

sensitivity models.

The contributing mass equations are given here:

( ) init

coolant

coolant

ins sc insinsins sc

F

FF

F GRC F

scscscsc

TTQ

cp m

d lm

d lm

alm

− =

τ



 



 τ

+πρ=

τ



 



 τ

−πρ=

ρ=

max

22

2

22

2

where

ρsc density of the superconducting wire (BSCCO) 

ρGRC density of the solenoid former (GRC) 

ρins density of the dielectric insulation (LPP)

τF solenoid former thickness

τins insulation thickness

5.6.9 Assumptions Overview

The SFCL voltage sensitivity study assumptions are provided in Table 26. SFCL sensitivity study 

variable ranges are shown in Table 27, and SFCL sensitivity study optimization variables are shown in 

Table 28. 
NASA/CR—2015-218440 125

TABLE 26.—SFCL VOLTAGE SENSITIVITY STUDY ASSUMPTIONS

Variable Value

Insulator dielectric strength (LPP), κins .....................................................................50 kV/mm

Final superconductor temperature, T..................................................................................92 K 

Coolant (starting) temperature, Ta0.....................................................................................70 K 

Superconductor critical temperature, Tc .............................................................................92 K 

Ambient final temperature after quench, Taf ......................................................................77 K 

Flux-creep region exponent (at 77K), n77K...............................................................................6 

Flux-flow region exponent, β...................................................................................................3 

Initial critical current density, Jc0 ......................................................................... 1.5×108 A/m2

Initial electric field, Ec ............................................................................................0.0001 V/m

Electric field at transition from flux-creep state to flux-flow state, E0 ..........................0.1 V/m

Normal conducting state resistivity (at Tc), ρ ......................................................0.000001 Ωm

Coefficient for heat transfer to cooling reservoir, κ .............................................1500 W/K m2

Superconductor volumetric specific heat, cv ....................................................1000000 J/K m3

Superconductor density, ρsc..................................................................................... 6500 kg/m3

Max magnetic flux density, B ........................................................................................... 0.5 T 

Dielectric Insulation density, ρinsulation ....................................................................... 900 kg/m3

Former density, ρformer ............................................................................................. 1800 kg/m3

Coolant medium.................................................................................................................... N2

Former thickness, τF......................................................................................................... 6 mm

Coolant (LN2), cp, ...................................................................................2.05 at 77 K kJ/kg*K

Coolant density (LN2), ΡLN2 ...................................................................................... 810 kg/m3

TABLE 27.—SFCL SENSITIVITY STUDY VARIABLE RANGES

Variable Values

Power (MW).................................................................................................... 11.2

Voltage (kV) (RMS if AC, ±X if DC)...................2, 4, 6, 8, 10, 12.5, 15, 17.5, 20

Fault current ratio......................................................................................... 2, 4, 6

Time to isolation (μs) ............................................................... 100, 5000, 100000

Min desired inductance (μH)...................................................... 10, 1000, 100000

Max allowable inductance (μH) ........................................................Not specified

Min quench resistance (Ω) ..................................................................... 0.1, 0.5, 1

Time to heat dissipation (s) ...................................................................... 1, 10, 30

TABLE 28.—SFCL SENSITIVITY STUDY OPTIMIZATION VARIABLES

Variable Range

Initial cryogenic temperature (K) Depends of coolant

Number of turns [1,∞)

Diameter of solenoid (m) [0.01,100]

Limit Range

Min quench resistance (Ω) See variable ranges

L/d of solenoid [0.1,100]

Inductance (μH) See variable ranges
NASA/CR—2015-218440 126

5.6.9.1 Mass Trends

In general, results show that increasing the operating voltage has favorable effects on the mass of the 

system. The magnitude of these effects depends on the protection and geometric requirements placed on 

the device. 

The sensitivity of the SFCL mass to operating voltage and other protection parameters (overcurrent, 

desired inductance, and time to isolation) is shown in Figure 128, Figure 129, and Figure 130. 

The ratio of fault current to nominal operating current primarily effects the SFCL mass via three 

physical properties: cryogenic fluid reservoir volume (variation in time to quench and energy dissipated), 

and solenoid winding (maximum magnetic field). As illustrated in Figure 128, increasing the fault current 

ratio has a proportional effect on the SFCL mass. It also has a significant impact on the degree to which 

voltage effects mass. 

With the variable settings indicated in the figure, for a ratio of fault to nominal current ratio of 9 the 

mass of the SFCL varies from 16 kg at 2 kV to 2 kg at 40 kV. Whereas, for a fault current to nominal 

current ratio of 3, the mass varies from approximately 6 to 2 kg on the same range. 

Figure 128.—SFCL Sensitivity of Mass to Voltage with Varying Fault Current Ratios. 

Figure 129.—SFCL Sensitivity of Mass to Voltage with Varying Inductance Requirements.
NASA/CR—2015-218440 127

In desired inductance has a similar impact on the SRCL mass trends. As shown in Figure 129, a 

higher desired inductance requires a larger device and more sensitivity to voltage. The manner in which 

the mass inductance requirements impact the weight is not directly through the field and cryogenic system 

mass. The required inductance effects mass directly through impacts to the solenoid geometry (number of 

windings, turns, and diameter). 

The time to isolation requirement has a singular impact on the mass of the SFCL. This value only impacts 

the mass and volume of the cryocooling fluid reservoir. The impact of this mass is shown in Figure 130. 

The benefit of a fault current limiter is its ability to increase resistance during a fault. The level of 

resistance required after quenching affects the overall system mass as illustrated in Figure 131. This mass 

increase is due to the increased heat generated during the fault and the increased length of the superconductor. 

SFCL performance also affects the cryogenic system mass. The amount of cooling power required 

from the cryocooler depends on the time in which the heat generated during a quench/isolation event to be 

dissipated. These mass trends are indicated in Figure 132. 

The voltage trends for all of SFCL components are shown in Figure 133 and Figure 134. The weight 

of all components goes down with voltage with the exception of the dielectric insulation. 

Additionally, the mass of the overall SFCL assembly is sensitive to the required impedance. Increased 

quench resistance and inductance can dramatically increase the SFCL weight.

Figure 130.—SFCL Sensitivity of Mass to Voltage with Varying Breaker Response Times.

Figure 131.—SFCL Sensitivity of Mass to Voltage with Varying Minimum Required 

Quench Resistance.
NASA/CR—2015-218440 128

Figure 132.—SFCL Sensitivity of Mass to Voltage with Varying Time to Dissipate 

Quench Energy with Cryogenic Systems. 

Figure 133.—Mass Contributions for Superconducting Fault Current Limiter for 

with Low Quench Resistance and Low Inductance Requirements.

Figure 134.—Mass Contributions for Superconducting Fault Current Limiter for 

with Higher Quench Resistance and Higher Inductance Requirements.
NASA/CR—2015-218440 129

5.6.10 Geometry Sensitivity

Geometry limits play a large role in limiting the benefits available. Field constraints affect the mass of 

the SCFL at lower voltages and geometry constraints begin to act on the mass as voltage increases. This is 

illustrated in Figure 135, Figure 136, and Figure 137. 

In Figure 135 the magnitude of the max magnetic field density limit does not impact the SFCL mass 

at voltages over approximately 15 kV for the variables settings indicated on the graph. For voltages under 

15 kV, increasing in the max allowable magnetic field density produces decreases in SFCL mass. 

Both diameter constraints and L/D constraints drive the unit mass at higher voltages; depending on the 

requirements. This is illustrated in Figure 136 and Figure 137. 

As shown in Figure 136, for each Lf/df limit corresponds to a point at which increased voltage may 

not yield added mass improvements. For the min Lf/df case of 1 the voltage sensitivity curve flattens 

around 20 kV while for a min Lf/df of 10, this point of diminished returns occurs around 6 kV. 

A similar point of diminishing returns occurs when imposing a minimum diameter limit on the SFCL 

former. An increase in min diameter has a proportional effect on the mass at higher voltage. 

Figure 135.—SFCL Sensitivity of Mass to Voltage with Varying the Maximum 

Allowable Magnetic Flux Densities.

Figure 136.—SFCL Sensitivity of Mass to Voltage with Varying Minimum L/D Limits 

on the Solenoid Geometry.

0

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3

4

5

6

7

8

9

0 10 20 30 40 50

Total Mass (kg)

Voltage (kV)

Superconducting Fault Current Limiter Voltage Sensitivity

Max Field (T)=0.5

Max Field (T)=1

Max Field (T)=2

Fault Current Ratio = 6,Time to Isolation (s) = 0.00005,Desired Inductance (H) = 0.00001,Lf/df_min = 1, df_min (m) = 0.05

0

1

2

3

4

5

6

7

8

9

0 10 20 30 40 50

Total Mass (kg)

Voltage (kV)

Superconducting Fault Current Limiter Voltage Sensitivity

Lf/df_min=1

Lf/df_min=5

Lf/df_min=10

Fault Current Ratio = 6,Time to Isolation (s) = 0.000005,Desired Inductance (H) = 0.00001,Max Field (T) = 0.5,, df_min (m) = 
NASA/CR—2015-218440 130

Figure 137.—SFCL Sensitivity of Mass to Voltage with Varying Minimum Diameter 

Limits on the Solenoid Geometry.

Figure 138.—Effect of Geometry Limits and Increased Inductance on SFCL Mass 

Sensitivity of Mass to Voltage.

In some instances, geometric constraints create a clear optimal voltage level for the SFCL. If the 

winding spacing is held fixed by field and wire geometry and the inductance value is allowed to float 

above the specified minimum inductance, geometric constraints may increase the SFCL mass as the 

voltage increases. In these scenarios, the decreased voltage impacts the winding spacing due to reduction 

in wire diameter. Therefore, this yields an increase in the number of windings available on the fixed 

geometry solenoid. In turn, the number of windings increases, which drives the inductance up. This is 

illustrated in Figure 138, Figure 139, and Figure 140. 

0

1

2

3

4

5

6

7

8

9

0 10 20 30 40 50

Total Mass (kg)

Voltage (kV)

Superconducting Fault Current Limiter Voltage Sensitivity

df_min (m)=0.05

df_min (m)=0.1

df_min (m)=0.15

Fault Current Ratio = 6,Time to Isolation (s) = 0.000005,Desired Inductance (H) = 0.00001,Max Field (T) = 0.5,Lf/df_min = 5

0

1

2

3

4

5

6

7

8

9

0 10 20 30 40 50

Total Mass (kg)

Voltage (kV)

Superconducting Fault Current Limiter Voltage Sensitivity

Lf/df_min=1

Lf/df_min=5

Lf/df_min=10

Fault Current Ratio = 9,Time to Isolation (s) = 0.0005,Desired Inductance (H) = 0.000001,Max Field (T) = 1,, df_min (m) = 0.05
NASA/CR—2015-218440 131

Figure 139.—Increased Number of Windings with Increased Voltage.

Figure 140.—Increase in Inductance at Higher Voltages due to Geometry Limits.

The increase in mass for the higher L/d limited cased in Figure 138 is a result of the increased number 

of windings (shown in Figure 139), with the increase in inductance from the increase in induction (shown 

in Figure 140).

5.6.11 Efficiency Trends

The efficiency of the SFCL is a function of the length of the superconducting wire and the 

superconducting resistive losses per unit wire length. 

sc sc l

P

EV R =

where

Rsc minimum required inductance

E per unit length superconducting electric field 

P power required

lsc length of the superconducting wire 

0

20

40

60

80

100

120

140

160

180

200

0 10 20 30 40 50

Number of turns

Voltage (kV)

Superconducting Fault Current Limiter Voltage Sensitivity

Lf/df_min=1

Lf/df_min=5

Lf/df_min=10

Fault Current Ratio = 6,Time to Isolation (s) = 0.000005,Desired Inductance (H) = 0.00001,Max Field (T) = 1,, df_min (m) = 

0

20

40

60

80

100

120

140

160

180

0 10 20 30 40 50

Inductance (

μH)

Voltage (kV)

Superconducting Fault Current Limiter Voltage Sensitivity

Lf/df_min=1

Lf/df_min=5

Lf/df_min=10

Fault Current Ratio = 6,Time to Isolation (s) = 0.00005,Desired Inductance (H) = 0.00001,Max Field (T) = 1,, df_min (m) = 0.05
NASA/CR—2015-218440 132

It is assumed that the power required remains constant at the design point, and that the electric field 

losses are defined for the critical current density at this design point. The resistance is directly 

proportional to the design voltage and the length of the superconducting wire (plotted against each other 

in Figure 141. 

Resistance trends are illustrated in Figure 142 to Figure 145. In general, it is observed that as the 

design voltage increases the length of the superconducting cable required decreases. However, the critical 

current also decreases. Assuming a constant electric field loss per unit length at the critical current yields 

this upward trend. 

Figure 141.—Decreased Length of Superconducting Cable with Increased Voltage.

Figure 142.—Increase in SFCL Resistance w.r.t. Design Voltage for Varying Fault 

Current Ratio Requirements.
NASA/CR—2015-218440 133

Figure 143.—Increase in SFCL Resistance w.r.t. Design Voltage for Varying the 

Time to Isolation.

Figure 144.—Increase in SFCL Resistance w.r.t. Design Voltage for Varying 

Solenoid Geometry Limits.

Figure 145.—Increase in SFCL Resistance w.r.t. Design Voltage for Varying Desired 

Quench Resistance.
NASA/CR—2015-218440 134

Figure 146.—Parameter Diagram for Motor Generator Model.

5.7 Generators

Generator weight and efficiency estimates were calculated using a high temperature superconducting 

machine design tool provided by NASA (Ref. 171). The weight and efficiency sensitivity assessments 

follow the parameter diagram in Figure 146. 

The primary parameters used in machine scaling are rotor speed, AC frequency, and peak power 

rating. A fixed topology and superconducting type are assumed in this sensitivity. A three pole BSCCO 

square wire wound generator is assumed in this tool’s machine sizing and performance sensitivity.

5.8 Cryogenic System 

Very rough estimates of the weight penalty associated with system inefficiencies were used in this 

study. The cryocooler system was assumed to be a 30 percent efficient reverse Brayton cycle system with 

a power density of 3 kg/kW. 

Cooling reservoirs, tubing, and flow management systems were not considered in detail for these 

parametric trade studies. 

5.9 Component Sensitivity Modeling Summary

Through this section, the governing equations and development of each component model were 

presented along with the resulting component mass and efficiency trends as a function of the DC 

distribution voltage. Each individual component (generator, power converter, cable, circuit breaker, SFCL, 

and SMES) has an optimum mass or efficiency with corresponding DC bus voltage at which that optimum 

is achieved. The next phase of this contract will include assembling all of the component models and 

corresponding weight and efficiency sensitivities to DC bus voltage to form the designated TeDP 

architecture selected for this study. The framework for this assembly and system sensitivity modeling is 

discussed in Section 6.0. The system sensitivity model will be used to understand the system mass and 

efficiency sensitivity to DC distribution voltage. It is expected that the mass contribution of each component 

will serve as a weighting function and influence how each component’s mass and efficiency sensitivity 

affect the system’s sensitivities. To determine the system sensitivities, the system model will be exercised 

by sweeping the DC distribution bus voltage and computing the system mass and efficiency trends for many 

component control parameters such as AC system frequency, temperature, and superconductor type. Based 

on the system sensitivities, a narrower range of DC bus voltages will be recommended. 

Generator/Motor

Weight

Efficiency

Heat Quality

Peak Rated 

Power

Topology

Rotor Speed Inertia Temperature Superconductor Type

AC Frequency 
NASA/CR—2015-218440 135

6.0 Narrowed DC Voltage Range

6.1 Introduction

The voltage range was narrowed by integrating the component sensitivity models generated for the prior 

deliverables. These models include sizing estimates for power electronics (converter, rectifiers), SMES energy 

(including converter), cables, superconducting fault current limiters, solid state circuit breakers, and electric 

machines. The models used for this systems sizing were described in the previous contract deliverables.

This integrated model was sampled over a range of operating voltages under series of assumptions 

regarding requirements and performance parameters. Additionally, the mass and power required for 

cryocooling was also captured by recording the cooling required for each system device, assuming a 

30 percent reverse Brayton cycle efficiency, and assuming a specific power of 5 hp/lb for the cryocooling 

equipment.

On- and off-design scenarios were identified which size the electrical equipment, the cryocoolers, and 

determine the nominal system efficiency. Considering fail-safe operations, the equipment sizing requirements 

are generated by the single-engine-out scenario at takeoff. A thrust power of 25 MW is required from one 

turbogenerator during this scenario. While these requirements size the electrical and cryogenic equipment, the 

nominal efficiency was calculated assuming a 10 MW cruise thrust power required. 

The energy storage system’s role was limited to assisting with temporary fill-in power during a source 

failure. 

The trends presented reflect general assumptions regarding protection system requirements. More 

accurate transient performance requirements for these devices will be explored as dynamic models of the 

system are completed and exercised. 

6.2 Architecture Assumptions

The system considered for voltage selection is outlined in Table 29. The number of each component 

is given as well as the nominal and peak power requirements.

TABLE 29.—ARCHITECTURE COMPONENT BREAKDOWN

Count Single engine out rating at takeoff,

MW

Nominal rating at cruise,

MW

Electric 

Machines

Generator 4 12.5 6.25

Motor 16 1.79 1.5625

Converter

AC/DC converter 4 12.5 6.25

DC/AC inverter 16 1.79 1.5625

DC/DC converter for SMES 4 12.5 0

Cables

AC 4 12.5 6.25

16 1.79 1.5625

Transmission 4 12.5 (2- by 30-m, 2- by 40-m) 6.25

Feeder 16 1.79 (16- by 5-m) 1.5625

16 1.34 (16- by 5-m) 0

Breakers

AC 12 12.5 6.25

48 1.79 1.5625

DC 16 12.5 6.25

64 1.79 1.5625

64 1.34 0

2 12.5 0

SFCL

AC 12 12.5 6.25

48 1.79 1.5625

DC 8 12.5 6.25

32 1.79 1.5625

32 1.34 0

2 12.5 0

En. storage SMES 4 12.5 0
NASA/CR—2015-218440 136

In the case of solid state circuit breakers and fault current limiters, the number of devices is large due 

to the number of poles or phases required for bi-polar DC and three-phase AC distribution. Each 

transmission and feeder line is configured to have two breakers at each end of the run to isolate line 

failures. Therefore, there are a total of four breakers per DC cable run. There is a single pair of fault 

current limiters per DC cable run and a set of three for each AC run.

Baseline assumptions for mass results displayed here are outlined in Table 30. 

TABLE 30.—SYSTEM MODEL PARAMETERS BASELINE ASSUMPTIONS

Generator

Turbine speed ................................................................8000 rpm

Pole count................................................................................... 3 

Fault current ratio ....................................................................... 2 

Motor

Propulsor speed ............................................................4000 rpm

Pole count................................................................................... 3 

Fault current ratio ....................................................................... 2 

Cables

Transmission lengths............................................................ 30 m 

.............................................................................................. 40 m 

AC lengths.............................................................................. 1 m

Feeder length.......................................................................... 5 m

Temperature rise on cable run ............................................. 0.5 K 

Fault current ratio ....................................................................... 2 

SFCL

Desired inductance .......................................................... 100 μ H

Desired quench resistance ...................................................... 1 Ω

Inductor core flux density......................................................0.5T

Time to heat dissipation w/ fault ...............................................5s

Fault current ratio ....................................................................... 2 

SMES

Stored energy (4 s at 12.5 MW) ......................................... 50 MJ

SMES voltage ratio to bus voltage ........................................ 0.25

Temperature ............................................................................. 18

Major to minor radius ratio......................................................... 4 

Compressive quality factor...................................................... 0.5

Temperature rise across SMES............................................ 0.5 K 

Converters 

Fault current ratio ....................................................................... 2 

Temperature ....................................................................... 100 K 

Modulation index ....................................................................... 1 

Power factor ............................................................................... 1 

DC ripple voltage magnitude.................................................20%

DC ripple current magnitude .................................................10%

SSCB

Fault current ratio ....................................................................... 2 

Temperature ....................................................................... 100 K 
NASA/CR—2015-218440 137

6.3 Electrical System Mass Sensitivity to Voltage 

Assuming this configuration of system protection, the protection devices are the single largest 

contributing technology group to overall mass. This weight breakdown is illustrated in Figure 147. For 

this baseline architecture the optimal voltage is roughly ±4.5 kV. 

Reductions in system weight may be possible by balancing the current blocking capability of the 

breakers and the converters. The switches sized for the generator and propulsor power electronics were 

selected due to their ability to block fault currents. If the conversion devices prove sufficient in fault 

current interruption, additional SSCB devices can be removed. However, solid state switches may still be 

required to reroute power from the alternative sources during a failure scenario.

The baseline configuration uses SSCBs and SFCLs for the protection of each device within the 

system. The result is a protection strategy which includes 7 protection zones between a single generator 

and a single propulsor. This zonal configuration is illustrated in Figure 148 with solid blue lines on the L1 

brand. This protection strategy requires that each all of the devices highlighted in green and yellow be 

engaged in system protection. The alternative extreme would be to rely completely on the converter 

devices to provide fault isolation and only retain the SSCBs which allow for power rerouting during a 

failure. This eliminates all the yellow highlighted SSCBs in Figure 148 and results in a 3 zone protection 

scheme. This means that any feeder fault will result in a loss of the branch. However, it may remove the 

12 higher power AC SSCBs on the generator side and 48 lower power AC SSCBs on the motor side. 

Additionally, 16 high power DC SSCB are eliminated and 64 low power DC SSCB on the feeders. 

Figure 149 illustrates the weight breakdown for the system for this reduced protection equipment 

approach. Removal of all of these overly-protective devices results in a reduction in overall system weight 

of approximately 20 percent. 

Figure 147.—Weight Decomposition with Component Count from Table 1 and Assumptions from Table 2. 

0

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15000

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25000

1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 6.5 7 7.5 8 8.5 9 9.5 10

Mass (kg)

Voltage (±kV)

Bus Tie SSCB Mass (kg)

Secondary Feeder SSCB Mass (kg)

Primary Feeder SSCB Mass (kg)

Transmission SSCB Mass (kg)

Propulsor Side AC SSCB Mass (kg)

Gen Side AC SSCB Mass (kg)

SMES DC-DC Converter Mass (kg)

SMES Mass (kg)

Propulsor Power Electronics Mass (kg)

Generator Power Electronics Mass (kg)

Bus Tie SFCL Mass (kg)

Secondary Feeder SFCL Mass (kg)

Primary Feeder SFCL Mass (kg)

Transmission SFCL Mass (kg)

Propulsor Side AC SFCL Mass (kg)

Gen Side AC SFCL Mass (kg)

Propulsor Side AC Mass (kg)

Secondary Feeder Mass (kg)

Primary Feeder Mass (kg)

Gen Side AC Mass (kg)

Secondary Transmission Mass (kg)

Primary Transmission Mass (kg)

Motor Mass (kg)

Generator Mass (kg)
NASA/CR—2015-218440 138

Figure 148.—Zonal Protection Alternatives.

Figure 149.—Weight Decomposition with no Protection SSCBs or SFCLs. 

0

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25000

1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 6.5 7 7.5 8 8.5 9 9.5 10

Mass (kg)

Voltage (±kV)

Bus Tie SSCB Mass (kg)

Secondary Feeder SSCB Mass (kg)

Primary Feeder SSCB Mass (kg)

Transmission SSCB Mass (kg)

Propulsor Side AC SSCB Mass (kg)

Gen Side AC SSCB Mass (kg)

SMES DC-DC Converter Mass (kg)

SMES Mass (kg)

Propulsor Power Electronics Mass (kg)

Generator Power Electronics Mass (kg)

Bus Tie SFCL Mass (kg)

Secondary Feeder SFCL Mass (kg)

Primary Feeder SFCL Mass (kg)

Transmission SFCL Mass (kg)

Propulsor Side AC SFCL Mass (kg)

Gen Side AC SFCL Mass (kg)

Propulsor Side AC Mass (kg)

Secondary Feeder Mass (kg)

Primary Feeder Mass (kg)

Gen Side AC Mass (kg)

Secondary Transmission Mass (kg)

Primary Transmission Mass (kg)

Motor Mass (kg)

Generator Mass (kg)
NASA/CR—2015-218440 139

In addition to addressing specific technology improvements, other weight savings may be available 

by descoping or removal of the energy storage, eliminating fault current limiters, or refocusing on AC 

distribution with oversized AC lines to reduce conductivity losses. While this approach does reduce the 

ability to independently throttle the propulsors, it reduces the weight of the converters and eliminates 

converter switching and conduction losses. This benefit must be balanced with the increase the AC 

conduction losses in the distribution system.

It is likely that the actual protection and recovery solution will yield a system which sits between the 

baseline and minimalist systems considered in this section. Configuring the system for fault isolation and 

recovery depends on the results of dynamic models. However, evaluation of the voltage sensitivity for 

both of these systems indicates that the optimal voltage is not highly dependent on the number and ratings 

of the protection devices.

6.4 Electrical System Cooling Requirements Sensitivity to Voltage

Losses play a major role in determining the overall system mass. Figure 150 and Figure 151 illustrate 

the overall heat load to the cryo-cooling system generated during nominal and peak loading scenarios. 

The major contributors to the heat load which must be managed by the cryogenic cooling system can

be sources to the SSCBs (pink layers) and the converters (orange layers). The solid state circuit breaker 

losses decrease with voltage, while the propulsor converter losses increase with voltage. These trends 

follow the IGBT and diode device scaling discussed in the previous deliverable.

Figure 150.—Heat from Devices during 10 MW Nominal Operation for Baseline Configuration and Losses. 

0

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2500

1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 6.5 7 7.5 8 8.5 9 9.5 10

Heat Load (kW)

Voltage (±kV)

Bus Tie SSCB Heat Load for Nominal (kW)

Secondary Feeder SSCB Heat Load for Nominal (kW)

Primary Feeder SSCB Heat Load for Nominal (kW)

Transmission SSCB Heat Load for Nominal (kW)

Propulsor Side AC SSCB Heat Load for Nominal (kW)

Gen Side AC SSCB Heat Load for Nominal (kW)

SMES DC-DC Converter Heat Load for Nominal (kW)

SMES Heat Load for Nominal (kW)

Propulsor Power Electronics Heat Load for Nominal 

(kW)

Generator Power Electronics Heat Load for Nominal 

(kW)

SMES DC-DC Converter Heat Load for Nominal (kW)

Secondary Feeder SFCL Heat Load for Nominal (kW)

Primary Feeder SFCL Heat Load for Nominal (kW)

Transmission SFCL Heat Load for Nominal (kW)

Propulsor Side AC Heat Load for Nominal (kW)

Gen Side AC SFCL Heat Load for Nominal (kW)

Propulsor Side AC Heat Load for Nominal (kW)

Secondary Feeder SSCB Heat Load for Nominal (kW)

Primary Feeder Heat Load for Nominal (kW)

Gen Side AC Heat Load for Nominal (kW)

Secondary Transmission Heat Load for Nominal (kW)

Primary Transmission Heat Load for Nominal (kW)

Motor Heat Load for Nominal (kW)

Generator Heat Load for Nominal (kW)
NASA/CR—2015-218440 140

Figure 151.—Temporary Heat from Devices during 25 MW Single Engine Out at Takeoff 

Scenario for Baseline Configuration and Losses.

The minimum heat load from the baseline system during nominal operation is approximately 800 kW. 

This represents an 8 percent of the overall 10 MW nominal load. Assuming a 30 percent efficient 

cryocooler and the operating temperatures in Table 2, the power required for cryocooling is a little less 

than 6 MW. 

For the single engine out scenario at takeoff, the power required is 25 MW. Fill-in power is provided 

temporarily by the energy storage (illustrated in Figure 151). After this support during system 

reconfiguration, the loss sensitivity to voltage follows the trends in Figure 152. For this baseline 

configuration, the steady state losses during this scenario are equal to roughly 9 percent of the overall 

power. For this magnitude of losses, the overall cryocooling power required is roughly 16 MW. 

As with mass reduction, it is expected that the losses will be reduced by reducing the scope of the 

protection equipment. Removing all SSCB which are not allocated to rerouting of power yields 

improvements illustrated in Figure 152 and Figure 153 (this does not capture the increase in size and 

losses due to modifications to converter requirements). Conversion losses and rerouting SSCB’s are 

retained for these trends. 

Comparing the results illustrated in Figure 149, Figure 150, and Figure 151 with those illustrated in 

Figure 152, Figure 153, and Figure 154, there is an obvious reduction in overall heat loss to the cryogenic 

system with a reduction in protection equipment. Also, large low voltage losses associated with the 

SSCB’s is also largely mitigated and a clear optimal voltage which minimized the losses is evident.

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3000

4000

5000

6000

1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 6.5 7 7.5 8 8.5 9 9.5 10

Heat Load (kW)

Voltage (±kV)

Bus Tie SSCB Heat Load for SEI@TO (kW)

Secondary Feeder SSCB Heat Load for SEI@TO (kW)

Primary Feeder SSCB Heat Load for SEI@TO (kW)

Transmission SSCB Heat Load for SEI@TO (kW)

Propulsor Side AC SSCB Heat Load for SEI@TO (kW)

Gen Side AC SSCB Heat Load for SEI@TO (kW)

SMES DC-DC Converter Heat Load for SEI@TO (kW)

SMES Heat Load for SEI@TO (kW)

Propulsor Power Electronics Heat Load for SEI@TO

(kW)

Generator Power Electronics Heat Load for SEI@TO

(kW)

Bus Tie SFCL Heat Load for SEI@TO (kW)

Secondary Feeder SFCL Heat Load for SEI@TO (kW)

Primary Feeder SFCL Heat Load for SEI@TO (kW)

Transmission SFCL Heat Load for SEI@TO (kW)

Propulsor Side AC SFCL Heat Load for SEI@TO (kW)

Gen Side AC SFCL Heat Load for SEI@TO (kW)

Propulsor Side AC Heat Load for SEI@TO (kW)

Secondary Feeder Heat Load for SEI@TO (kW)

Primary Feeder Heat Load for SEI@TO (kW)

Gen Side AC Heat Load for SEI@TO (kW)

Secondary Transmission Heat Load for SEI@TO (kW)

Primary Transmission Heat Load for SEI@TO (kW)

Motor Heat Load for SEI@TO (kW)

Generator Heat Load for SEI@TO (kW)
NASA/CR—2015-218440 141

Figure 152.—Heat from Devices during 10 MW Nominal Operation for Configuration with no 

Protection SSCB’s or SFCL’s. 

Figure 153.—Temporary Heat from Devices during 25 MW Single Engine Out at Takeoff 

Scenario for Configuration with no Protection SSCB’s or SFCL’s. 

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600

700

800

900

1000

1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 6.5 7 7.5 8 8.5 9 9.5 10

Heat Load (kW)

Voltage (±kV)

Bus Tie SSCB Heat Load for Nominal (kW)

Secondary Feeder SSCB Heat Load for Nominal (kW)

Primary Feeder SSCB Heat Load for Nominal (kW)

Transmission SSCB Heat Load for Nominal (kW)

Propulsor Side AC SSCB Heat Load for Nominal (kW)

Gen Side AC SSCB Heat Load for Nominal (kW)

SMES DC-DC Converter Heat Load for Nominal (kW)

SMES Heat Load for Nominal (kW)

Propulsor Power Electronics Heat Load for Nominal

(kW)

Generator Power Electronics Heat Load for Nominal

(kW)

SMES DC-DC Converter Heat Load for Nominal (kW)

Secondary Feeder SFCL Heat Load for Nominal (kW)

Primary Feeder SFCL Heat Load for Nominal (kW)

Transmission SFCL Heat Load for Nominal (kW)

Propulsor Side AC Heat Load for Nominal (kW)

Gen Side AC SFCL Heat Load for Nominal (kW)

Propulsor Side AC Heat Load for Nominal (kW)

Secondary Feeder SSCB Heat Load for Nominal (kW)

Primary Feeder Heat Load for Nominal (kW)

Gen Side AC Heat Load for Nominal (kW)

Secondary Transmission Heat Load for Nominal (kW)

Primary Transmission Heat Load for Nominal (kW)

Motor Heat Load for Nominal (kW)

Generator Heat Load for Nominal (kW)

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500

1000

1500

2000

2500

3000

1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 6.5 7 7.5 8 8.5 9 9.5 10

Heat Load (kW)

Voltage (±kV)

Bus Tie SSCB Heat Load for SEI@TO (kW)

Secondary Feeder SSCB Heat Load for SEI@TO (kW)

Primary Feeder SSCB Heat Load for SEI@TO (kW)

Transmission SSCB Heat Load for SEI@TO (kW)

Propulsor Side AC SSCB Heat Load for SEI@TO (kW)

Gen Side AC SSCB Heat Load for SEI@TO (kW)

SMES DC-DC Converter Heat Load for SEI@TO (kW)

SMES Heat Load for SEI@TO (kW)

Propulsor Power Electronics Heat Load for SEI@TO

(kW)

Generator Power Electronics Heat Load for SEI@TO

(kW)

Bus Tie SFCL Heat Load for SEI@TO (kW)

Secondary Feeder SFCL Heat Load for SEI@TO (kW)

Primary Feeder SFCL Heat Load for SEI@TO (kW)

Transmission SFCL Heat Load for SEI@TO (kW)

Propulsor Side AC SFCL Heat Load for SEI@TO (kW)

Gen Side AC SFCL Heat Load for SEI@TO (kW)

Propulsor Side AC Heat Load for SEI@TO (kW)

Secondary Feeder Heat Load for SEI@TO (kW)

Primary Feeder Heat Load for SEI@TO (kW)

Gen Side AC Heat Load for SEI@TO (kW)

Secondary Transmission Heat Load for SEI@TO (kW)

Primary Transmission Heat Load for SEI@TO (kW)

Motor Heat Load for SEI@TO (kW)

Generator Heat Load for SEI@TO (kW)
NASA/CR—2015-218440 142

Figure 154.—Steady State Heat from Devices during 25 MW Single Engine Out at Takeoff 

Scenario for Baseline Configuration.

6.5 IGBT and Diode Switching and Conduction Losses

An alternate method to descoping the protection system is to target device improvements. The largest 

contributor to system inefficiencies are the switching and conduction losses in the converters and SSCB’s 

generated by the IGBT’s and diodes. Improvement in switching and conduction efficiency has a dramatic 

effect on the overall system weight. Improvements in IGBT and diode efficiency of 50 and 90 percent are 

illustrated in Figure 154, Figure 155, and Figure 156. 

The improvements illustrated in these figures also drive corresponding decreases in cryocooler power 

requirements.

6.6 Cryocooler Mass 

Depending on the redundancy of the cryogenic cooling systems, its mass lies between two limits 

illustrated in Figure 157. The lower limit is generated by the total heat load during the takeoff scenario. 

This scenario requires the largest magnitude of thrust power and the largest magnitude of heat rejection to 

the thermal management system. The upper limit is a system with components dedicated to each electrical 

device exclusively as sized for the single engine out at takeoff scenario. 

Cryogenic system mass reductions are available with reduction in overall heat rejection as discussed 

in the previous section. Reductions in the mass range for these systems are illustrated in Figure 158. 

As is evident from these trends, the optimal mass of the system can be more a function of the 

efficiency of the electrical system than the mass of the system itself. Mass improvements on the electrical 

side are desirable. However, these improvements must avoid associated increases in heat loss to the cryosystem. Figure 159 illustrates the relative size of the cryo-system (in blue) to the electric system (in 

purple) with variations in the switching and conduction efficiency. 

0

1000

2000

3000

4000

5000

6000

1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 6.5 7 7.5 8 8.5 9 9.5 10

Heat Load (kW)

Voltage (±kV)

Bus Tie SSCB Heat Load for SEI@TO (kW)

Secondary Feeder SSCB Heat Load for SEI@TO (kW)

Primary Feeder SSCB Heat Load for SEI@TO (kW)

Transmission SSCB Heat Load for SEI@TO (kW)

Propulsor Side AC SSCB Heat Load for SEI@TO (kW)

Gen Side AC SSCB Heat Load for SEI@TO (kW)

SMES DC-DC Converter Heat Load for SEI@TO (kW)

SMES Heat Load for SEI@TO (kW)

Propulsor Power Electronics Heat Load for SEI@TO

(kW)

Generator Power Electronics Heat Load for SEI@TO

(kW)

Bus Tie SFCL Heat Load for SEI@TO (kW)

Secondary Feeder SFCL Heat Load for SEI@TO (kW)

Primary Feeder SFCL Heat Load for SEI@TO (kW)

Transmission SFCL Heat Load for SEI@TO (kW)

Propulsor Side AC SFCL Heat Load for SEI@TO (kW)

Gen Side AC SFCL Heat Load for SEI@TO (kW)

Propulsor Side AC Heat Load for SEI@TO (kW)

Secondary Feeder Heat Load for SEI@TO (kW)

Primary Feeder Heat Load for SEI@TO (kW)

Gen Side AC Heat Load for SEI@TO (kW)

Secondary Transmission Heat Load for SEI@TO (kW)

Primary Transmission Heat Load for SEI@TO (kW)

Motor Heat Load for SEI@TO (kW)

Generator Heat Load for SEI@TO (kW)
NASA/CR—2015-218440 143

Figure 155.—Steady State Heat from Devices during 25 MW Single Engine Out at Takeoff 

Scenario for Baseline Configuration with 50 percent Decrease in Switching and Conduction 

Losses.

Figure 156.—Steady State Heat from Devices during 25 MW Single Engine Out at Takeoff 

Scenario for Baseline Configuration with 90 percent Decrease in Switching and Conduction 

Losses.

0

500

1000

1500

2000

2500

3000

3500

1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 6.5 7 7.5 8 8.5 9 9.5 10

Heat Load (kW)

Voltage (±kV)

Bus Tie SSCB Heat Load for SEI@TO (kW)

Secondary Feeder SSCB Heat Load for SEI@TO (kW)

Primary Feeder SSCB Heat Load for SEI@TO (kW)

Transmission SSCB Heat Load for SEI@TO (kW)

Propulsor Side AC SSCB Heat Load for SEI@TO (kW)

Gen Side AC SSCB Heat Load for SEI@TO (kW)

SMES DC-DC Converter Heat Load for SEI@TO (kW)

SMES Heat Load for SEI@TO (kW)

Propulsor Power Electronics Heat Load for SEI@TO

(kW)

Generator Power Electronics Heat Load for SEI@TO

(kW)

Bus Tie SFCL Heat Load for SEI@TO (kW)

Secondary Feeder SFCL Heat Load for SEI@TO (kW)

Primary Feeder SFCL Heat Load for SEI@TO (kW)

Transmission SFCL Heat Load for SEI@TO (kW)

Propulsor Side AC SFCL Heat Load for SEI@TO (kW)

Gen Side AC SFCL Heat Load for SEI@TO (kW)

Propulsor Side AC Heat Load for SEI@TO (kW)

Secondary Feeder Heat Load for SEI@TO (kW)

Primary Feeder Heat Load for SEI@TO (kW)

Gen Side AC Heat Load for SEI@TO (kW)

Secondary Transmission Heat Load for SEI@TO (kW)

Primary Transmission Heat Load for SEI@TO (kW)

Motor Heat Load for SEI@TO (kW)

Generator Heat Load for SEI@TO (kW)

0

100

200

300

400

500

600

700

800

900

1000

1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 6.5 7 7.5 8 8.5 9 9.5 10

Heat Load (kW)

Voltage (±kV)

Bus Tie SSCB Heat Load for SEI@TO (kW)

Secondary Feeder SSCB Heat Load for SEI@TO (kW)

Primary Feeder SSCB Heat Load for SEI@TO (kW)

Transmission SSCB Heat Load for SEI@TO (kW)

Propulsor Side AC SSCB Heat Load for SEI@TO (kW)

Gen Side AC SSCB Heat Load for SEI@TO (kW)

SMES DC-DC Converter Heat Load for SEI@TO (kW)

SMES Heat Load for SEI@TO (kW)

Propulsor Power Electronics Heat Load for SEI@TO

(kW)

Generator Power Electronics Heat Load for SEI@TO

(kW)

Bus Tie SFCL Heat Load for SEI@TO (kW)

Secondary Feeder SFCL Heat Load for SEI@TO (kW)

Primary Feeder SFCL Heat Load for SEI@TO (kW)

Transmission SFCL Heat Load for SEI@TO (kW)

Propulsor Side AC SFCL Heat Load for SEI@TO (kW)

Gen Side AC SFCL Heat Load for SEI@TO (kW)

Propulsor Side AC Heat Load for SEI@TO (kW)

Secondary Feeder Heat Load for SEI@TO (kW)

Primary Feeder Heat Load for SEI@TO (kW)

Gen Side AC Heat Load for SEI@TO (kW)

Secondary Transmission Heat Load for SEI@TO (kW)

Primary Transmission Heat Load for SEI@TO (kW)

Motor Heat Load for SEI@TO (kW)

Generator Heat Load for SEI@TO (kW)
NASA/CR—2015-218440 144

Figure 157.—Upper and Lower Limits for Cryocooler Size with a Reduction.

Figure 158.—Reductions in Cryogenic System Mass Due to Converter Efficiency Improvements.

6.7 Narrowed Voltage Range 

For all the configuration and converter efficiency variations of the architecture discussed here, the 

optimal voltage level never exceeds ±4.5 kV. With architecture changes and efficiency improvements, the 

optimal voltage may drop as low as ±2 kV. This remains consistent with variations in the converter 

efficiency (Table 31).

0

50000

100000

150000

200000

250000

1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 6.5 7 7.5 8 8.5 9 9.5 10

Mass (kg)

Voltage (±kV)

Total Mass for Independent

Coolers (kg)

Total Mass for Peak Heating

SEO@TOl (kg)
NASA/CR—2015-218440 145

TABLE 31.—OPTIMAL POLE VOLTAGE FOR A 25 MW TEDP SYSTEM

FOR VARIOUS CONFIGURATION AND EFFICIENCY IMPROVEMENTS

Baseline switching 

loss

50% improvement 

in converter losses

90% improvement 

in converter losses

Baseline system ±4.5 kV

[17,254 / 57,434]

±4.5 kV

[16,451 / 31,170]

±4.5 kV

[15,808 / 10,159]

Without protection SSCBs and all SFCLs ±3 kV

[11,816 / 33,659]

±3 kV

[11,126 / 18,872]

±4.5 kV

[10,828 / 6,611]

Without energy storage, protection SSCBs and SFCLs ±2 kV

[10,708 / 33,175]

±3 kV

[10,036 / 18,672]

±4.5 kV

[9.748 / 6,563]

Optimal Voltage (±kV) 

[Mass of electrical/cryo equipment (kg)]

A B C 

Figure 159.—Relative Mass of the Cryo and Electric Systems with Converter Efficiency Improvements 

(A- Baseline Switching and Conduction Losses, B- 50 percent Improvement, C- 90 percent 

Improvement).

Considering these results and taking into account both electrical and cooling contributions, the target 

voltage range for this system is between ±2 and ±4.5 kV. However, it is expected that system 

improvements necessary to reduce the losses to minimize cryogenic system mass will push this optimal 

voltage to the lower end of that range. The mass was also seen to be insensitive to voltage selection on the 

range of approximately ±2 kV. For a baseline system, a 2 kV increase or a 1 kV decrease from the sizing 

voltage only results in an approximate 5 percent increase in system mass. For the minimalist protection 

system, the target voltage range shifts as a function of converter efficiency. These 5 percent mass 

variability ranges are highlighted in Figure 160 and Figure 161. 

0

25000

50000

75000

100000

125000

150000

1 2 3 4 5 6 7 8 9 10

Mass (kg)

Voltage (±kV)

Total Mass for

Peak Heating

SEO@TOl (kg)

Total Mass for

Electrical

Equipment (kg)

0

25000

50000

75000

100000

125000

150000

1 2 3 4 5 6 7 8 9 10

Mass (kg)

Voltage (±kV)

Total Mass for

Peak Heating

SEO@TOl (kg)

Total Mass for

Electrical

Equipment (kg)

0

25000

50000

75000

100000

125000

150000

1 2 3 4 5 6 7 8 9 10

Mass (kg)

Voltage (±kV)

Total Mass for

Peak Heating

SEO@TOl (kg)

Total Mass for

Electrical

Equipment (kg)
NASA/CR—2015-218440 146

Figure 160.—Baseline Architecture Voltage Sensitivity.

Figure 161.—Reduced Protection System Architecture Voltage Sensitivity.
NASA/CR—2015-218440 147

7.0 GT Dynamic Model

7.1 Introduction

To identify regulation, protection, and recovery requirements, Georgia Tech/ASDL was tasked with 

developing preliminary models of switching, isolation, and protection components. The components that 

were modeled include: superconducting fault current limiter, solid-state circuit breaker, rectifier, inverter, 

and superconducting magnetic energy storage. Most of the models were created using the 

SimPowerSystems toolbox in MATLAB Simulink (The MathWorks, Inc.). Some state-space models were 

created as well in an effort to decrease the required time for simulation.

Once the protection component models were completed, dynamic system models were developed 

which captures the response of the state variables (current and voltage). Several failure scenarios were 

modeled including branch faults and engine failures. The current and voltage excursions observed in this 

system, as well as loss of propulsive power to the loads provide more accurate requirements to the TeDP 

system components and establish the voltage limits for normal and abnormal operating scenarios. 

7.2 Superconducting Fault Current Limiter

As discussed in the parametric sizing model development, superconducting fault current limiters can 

fall under two categories: resistive type and inductive type. The resistive type SFC has advantages of 

having a high limitation level and compact size (Ref. 172), so the resistive type was selected for the 

model. Resistive type SFCL can be made using either YBCO; however, Bi-2223 is commonly used since 

its resistivity changes faster with temperature (Ref. 173). Hence, Bi-2223 was selected for the models that 

will be described. The approach to identifying quench dynamics for this component follows similar 

assumptions as those applied for estimating the quench time for the parametric sizing model.

7.2.1 SFCL Modeling Overview

During the literature review, several approaches to modeling the resistance of the SFCL were 

available. The most widely used is the one that is presented in Blair and Nemdili (Refs. 173 and 174). 

With this approach, the resistance of the SFCL is determined based upon the operating point of the SFCL. 

The resistance of the SFCL is calculated as:

I

LE R sc SFCL

∗ =

Lsc is the length of the superconductor. I is the current flowing through the SFCL. E is the electric field. 

How the electric field is calculated is dependent on the state of the SFCL. The SFCL can operate in three 

regions: superconducting, flux flow, and normal conducting. The operating region of the SFCL can be 

determined using the SFCL temperature (Tsc) and electric field calculation from the previous iteration. 

(The initial electric field of the superconductor (E0) is an input provided by the user.) 

If the temperature of the SFCL is less than the superconductor critical temperature (Tc) and less than 

E0, then it is assumed that the SFCL is operating in the superconducting region. In this region, the electric 

field can be calculated as (Ref. 173): 

( )

T

TJ

J EE c

c

α















 =
NASA/CR—2015-218440 148

Ec is the critical electric field. J is the current density of the SFCL. Jc(T) is the temperature dependent 

critical current density of the superconductor. If the SFCL temperature is less than the superconductor 

temperature, the critical current density can be calculated as (Ref. 173): 

( ) c K

ac

c sc c J

TT

TT TJ  77 

 





−

− =

Ta is the ambient temperature. In the model, it is assumed that this is the temperature of LN2 (77 K). Jc77K

is the critical current density of the superconductor at the ambient temperature of 77K. If the temperature 

of the SFCL is higher than the critical temperature, then the critical current density is set to Jc77K. 

αT is a temperature dependent shaping parameter. It is calculated as (Ref. 173): 

[ ] ( )

( )

( )

( )





































































′ =α

β=α α′

β α − K

c c

c

c

T

E

E

TJ

J

E

E

T

T

K 77

1

0

1 1

0

77 log

log

max ,

β is another shaping parameter which is dependent on the material of the superconductor. α(77K) is the α 

shaping parameter at the ambient temperature of 77 K. This parameter is also dependent on the material 

selection. 

If the temperature of the SFCL is less than the critical temperature, but the electric field from the 

previous iteration is greater than the initial electric field, the SFCL is in the flux-flow region. In this 

region, the electric field is calculated as (Ref. 173): 

( ) ( )

( ) ( )

β α

β















































 = KJ

J

TJ

KJ

E

E EE c c

c K c

77

77 77

0

0

If both the critical temperature and the initial electric field are exceeded, then the SFCL is in the 

normal conduction region. In this region, the electric field is calculated as (Ref. 173): 

ρ= ( ) c JTE

ρ is the normal resistivity of the superconductor. Again, this value will be dependent on the material 

chosen for the superconductor. 

These calculations require that the temperature of the SFCL be calculated. The temperature of the 

SFCL will depend on the ambient temperature, resistance of the SFCL, current flow, and the heat transfer 

properties of the SFCL and coolant. 

7.2.2 SFCL SimPowerSystems Model

A model of the SFCL was created using SimPowerSystems. The model is shown in Figure 162. A 

simple power system was created to test the performance of the SFCL. The system has a source that 

consists of an ideal AC voltage source in series with a resistance and inductance. The system has a simple 

resistive load. A ground fault is placed in the system and activated at a time specified by the user of the 
NASA/CR—2015-218440 149

model. The resistance calculation block contains the algorithm for calculating SFCL resistance that was 

presented in the previous section. The temperature calculation is performed using a thermal equivalent 

circuit which is shown in Figure 163. The thermal model has an input of the SFCL current. Also, the 

thermal resistance and capacitance of the SFCL must be specified. In a thermal equivalent circuit, voltage 

is equivalent to temperature. So, the ambient temperature is represented by a voltage source is the system. 

The temperature of the SFCL is found by measuring the voltage across the capacitor in the circuit. 

The parameters that must be set by the user are shown in Table 32 along with the settings that were 

used for an example simulation. The large number of input variables allows the user to model a variety of 

types of SFCL designs.

An example simulation is presented to demonstrate the capabilities of the model. Of course, if input 

parameters are changed, (different system with a fault, different type of SCFL, etc.) the performance of 

the SFCL will change. 

Figure 162.—SFCL SimPowerSystems Model.

Figure 163.—SFCL Thermal Equivalent Circuit.
NASA/CR—2015-218440 150

TABLE 32.—SFCL SIMPOWERSYSTEMS MODEL PARAMETERS

Parameter Setting for example simulation

System parameters

Source voltage, V ................................................................................................110

Source frequency, Hz............................................................................................60

Line resistance, Ω ............................................................................................0.001

Line inductance, H .........................................................................................0.0006

Load resistance, Ω...................................................................................................2 

Fault start time, s ................................................................................................0.05

Duration of fault, s .............................................................................................0.01

Ambient temperature, K ........................................................................................77

Superconductor properties

Material density, kg/m3 ................................................................................ 9.2×103

Cross-sectional area, m2, (Ref. 175) ...........................................................2.4×10–6

Length, m..............................................................................................................22

Initial electric field, V/m, (Ref. 173) ....................................................................0.1

Critical electric field, V/m, (Ref. 173) ........................................................... 1×10–4

Critical current density at 77 K, A/m2, (Ref. 174)....................................... 1.5×107

Beta (Ref. 174) .......................................................................................................3 

Alpha at 77 K (Ref. 174).........................................................................................6 

Normal resistivity, Ω/m, (Ref. 174) ............................................................... 1×10–6

Critical temperature, K ..........................................................................................95

Thermal conductivity, W/(mK)............................................................................6.5

Heat capacity at constant pressure, J/gK, (Ref. 176) ........................................0.162

Figure 164.—System Fault Current without Protection.

First, the system was simulated without the SCFL to show the fault condition of the system without 

any protection. The fault current is shown in Figure 164, and the load voltage is shown in Figure 165. The 

results show a rapid rise in current when the fault occurs. The current during the fault rises to almost 10 

times the steady-state current. When the fault clears, there is a large increase in voltage across the load, 

and the voltage spikes to about 15 times the steady-state condition. 
NASA/CR—2015-218440 151

Figure 165.—Example System Load Voltage without Protection (V).

Figure 166.—Fault Current with SFCL (A)—SimPowerSystems Model.

Next, the simulation was performed with the added resistance from SFCL. The new fault current and 

load voltage are shown in Figure 166 and Figure 167. The change in resistance and temperature of the 

SFCL are shown in Figure 168 and Figure 169. The SFCL limits the current to about 400 A—smaller 

than half the magnitude of the fault current without any protection. Also, the magnitude of the voltage 

spike after the fault is greatly reduced. Rather than spiking to almost 1800 V, the voltage is limited to less 

than 300 V. These benefits are realized due to the increase in resistance of the SCFL. As shown in 

Figure 168, the resistance of the SCFL increases from almost zero to about 0.25 Ω. Figure 169 

demonstrates a temperature rise of about 3° in the SCFL. While this temperature rise will cause some 

changes in the SCFL, it is not enough to cause it to quench. Therefore, the increase in current must be due 

to an increase in electric field caused by the increase in current. Soon as the current is restored to normal 

levels, the SCFL will recover and return to a superconducting state.
NASA/CR—2015-218440 152

Figure 167.—Load Voltage with SFCL (V)—SimPowerSystems Model.

Figure 168.—SFCL Resistance (Ω)—SimPowerSystems Model.

Figure 169.—SFCL Temperature (K)—SimPowerSystems Model.
NASA/CR—2015-218440 153

7.2.3 SFCL State-Space Model

While the SimPowerSystems model represents the system well, it can be slow to run and also requires 

the use of the SimPowerSystems toolbox. To overcome these obstacles, a state-space model of the SFCL 

was also constructed. The state-space model solves a set of partial differential equations to find the 

response of the system. The set of equations can be found by using Kirchhoff’s circuit laws. Applying 

Kirchhoff’s voltage laws to the circuit shown in the SimPowerSystems model, the following three 

equations are obtained: 

( )

LLFF

s SFCL FF

RiRi

IR Ri

dt

dI tV IR L

=

+++=

s( ) IRSFCL Ri LL dt

dI tV IR L +++=

Vs is the source voltage. R is the line resistance. L is the line inductance. I is the current through the SFCL. 

RSFCL is the resistance of the SFCL. Rf is the resistance of the switch that causes the fault. Under normal 

conditions, this is set to a very high value. When the fault occurs, this resistance becomes very small. iF is 

the current through the switch causing the fault. iL is the current through the load. RL is the resistance of 

the load. 

Also, using Kirchhoff’s current law: 

+= iiI LF

Using the current law relation and the second voltage law equation, the following relation is obtained:

FL

F

L RR

IR i

+ =

Substituting into the third voltage law equations, the following circuit equation is obtained: 

( ) L

FL

F s SFCL R

RR

IR IR

dt

dI tV IR L

+

+++=

The equation can be solved to obtain the fault current in the system. The load voltage can be calculated as 

the product of the load current and load resistance. The resistance of the superconductor was calculated 

the same way as described in the beginning of this section and used in the SimPowerSystems model. 

Instead of using a thermal equivalent circuit, a different approach was used to calculate the temperature of 

the SFCL. The equations used were (Ref. 177): 

( ) TAW

RIQ

WQ dt

dT C

sc

∆α=

=

−=

2

C is the thermal capacitance of the superconductor or shunt. T is the temperature of the superconductor or 

shunt. I is current. α is the heat transfer coefficient to the coolant (set to 5×103 W/m2

K in the model 
NASA/CR—2015-218440 154

(Ref. 178)). ∆T is the temperature difference between the superconductor or shunt and the coolant. A is 

the surface area of the contact between the SFCL and the coolant.

The same system was simulated using the state-space model that was simulated with the 

SimPowerSystems model. The load current and voltage are shown in Figure 170 and Figure 171. The 

SFCL resistance and temperature are shown in Figure 172 and Figure 173. The results obtained by the 

state-space model are similar to those found using the SimPowerSystems model. The largest discrepancy 

is the SFCL temperature calculation. Two different models were used to calculate temperature. Further 

study is needed to determine the accuracy of each temperature model. The difference in the temperatures 

causes a slight difference in the calculated resistance of the SFCL (Ref. 179). This led to a small 

difference in the calculated fault current and load voltage. 

7.3 Solid-State Circuit Breaker

Solid-state circuit breakers use modern high-power semiconductors rather than electromechanical 

devices to protect a system from overload or a short circuit. Solid-state circuit breakers can react in a few 

microseconds versus a mechanical device that can take at least 100 ms to clear. With the fast response, the 

magnitude of the fault current will be greatly reduced and will help prevent damage to equipment in the 

system and prevent superconducting elements in the turboelectric system from quenching (Ref. 180).

Figure 170.—Fault Current (A)—State-space Model.

Figure 171.—Load Voltage (V)—State-Space Model.
NASA/CR—2015-218440 155

Figure 172.—SFCL Resistance (Ω)—State-Space Model.

Figure 173.—SFCL Temperature (K)—State-Space Model.

7.3.1 SSCB Modeling Overview

The circuit breaker topology that was modeled is shown in Figure 174. The IGBTs are on when the 

system is under normal operating conditions. When an over-current is sensed, the circuit breaker will be 

tripped and the IGBTs will be turned off.

Determining when to turn on and off the IGBTs is critical to the effectiveness of the SSCB. The 

protection scheme needs to be sensitive enough to quickly trip the circuit breaker when a fault occurs so 

that the fault current is minimized. However, if the control is overly sensitive, the circuit breaker may trip 

during a temporary over-current condition. 

The most widely used control scheme for SSCB is inverse-time over-current protection. This scheme 

senses the ratio of the actual current to a set tripping current. When an over-current is detected, the 

breaker will be tripped after a given amount of delay is determined using inverse-time curves. The inverse 

time curve can be calculated using the following equation (Ref. 179): 

9

5*14

1

tripping

trip

−





























= + −

TD B

I

I

A

t P
NASA/CR—2015-218440 156

I is the sensed current. Itripping is the tripping current. A, B, P, and TD (time dial) are all shaping 

parameters. Their settings are determined based upon how sensitive the designer wants the control 

scheme to be. A variety of curve options exist. A list is provided in Table 33, and Figure 175 shows the 

curves with a time dial setting of 0.5. (The lower the time dial setting, the faster the circuit breaker will be 

tripped.) 

A timer is started at the instant that an over-current is sensed. When the amount of time counted by 

the timer exceeds the calculated ttrip time, the breaker will be tripped. At any point during this period, if 

the sensed current falls to less than Itripping, the timer can be reset until an over-current is encountered 

again. 

TABLE 33.—INVERSE-TIME CURVES (REF. 179) 

Inverse-time curve A B P

Normal inverse 0.0086 0.0185 0.02

Very inverse 2.855 0.0712 2 

Extremely inverse 6.407 0.025 2 

Short time inverse 0.00172 0.0037 0.02

Short time extremely inverse 1.281 0.005 2 

Figure 174.—SSCB Circuit Diagram.

Figure 175.—Inverse-Time Curves with Time Dial Setting of 0.5.
NASA/CR—2015-218440 157

7.3.2 SSCB SimPowerSystems Model

A simple DC circuit was used to simulate the SSCB. The model is shown in Figure 176. The system 

consists of an ideal DC voltage source, a line resistance and inductance, SSCB, line-to-line fault, and a 

resistive load. To create the SSCB, the SimPowerSystems models for a diode, IGBT, and surge arrestor 

were used. (The design of the SSCB uses a varistor, but the SimPowerSystems surge arrestor model can 

be used to simulate a varistor given the correct settings.) The IGBTs are controlled using the inverse-time 

over-current protection scheme that was described in the previous section. 

The model allows the user to set a number of parameters. The input variables are shown in Table 34 

along with their settings for an example simulation that was performed. The results of the simulation are 

shown in Figure 177 to Figure 180. First the system was simulated without the circuit breaker. Figure 178 

and Figure 179 show the response of the system under this condition. Figure 178 shows that the fault 

current reaches nearly 6000 A. Next, the system was simulated with the circuit breaker in place. 

Figure 179 shows the fault current response, and Figure 180 shows the voltage across the load. With the 

use of the SSCB, the fault current is limited to about 1/3 of the fault current when no protection is used. 

The responses show that there is a slight delay between the start of the fault at 0.2 s and the activation of 

the circuit breaker. This delay is about 0.051 s. The size of this delay can be altered by changing the 

inverse-time over-current protection parameters.

TABLE 34.—SSCB MODEL PARAMETERS 

Parameter Setting for example simulation

System Parameters

Source voltage, V......................................................................6000

Line resistance, Ω ...........................................................................1 

Line inductance, H.................................................................... 0.02

Load resistance, Ω ..........................................................................6 

Fault start time, s..........................................................................0.2

Fault end time, s ...........................................................................0.3

Overcurrent Protection Parameters

Inverse-time curve ...............................................Short inverse time

Time dial......................................................................................0.3

Tripping current, A ...................................................................2000

IGBT Parameters (Ref. 181) 

On resistance, Ω................................................................. 0.000625

On inductance, H ............................................................................0 

Forward voltage, V .........................................................................1 

Current 10% fall time, s .......................................................40×10–9

Current tail time, s................................................................45×10–9

Initial current, S ..............................................................................0 

Snubber resistance, Ω ............................................................. 1×105

Snubber capacitance, °F............................................................... inf

Diode Parameters (Refs. 182 and 183) 

On resistance, Ω.......................................................................0.001

On inductance, H ............................................................................0 

Forward voltage, V ....................................................................0.85

Initial current, A..............................................................................0 

Snubber resistance, Ω ............................................................. 1×105

Snubber capacitance, F ......................................................250×10–9
NASA/CR—2015-218440 158

Figure 176.—SSCB Model.

Figure 177.—Example System Fault Current without Protection (A).

Figure 178.—Example System Load Voltage without Protection (V).
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Figure 179.—Fault Current with SSCB.

Figure 180.—Load Voltage with SSCB (V).

7.4 Energy Storage Model (SMES)

The energy storage device selected for this system is superconducting magnetic energy storage. The 

topology of the device is shown in Figure 181. 

Figure 181 shows that the SMES operates in one of three states: charging, steady-state, and discharging. 

The state of the SMES can be changed by switching the IGBTs in the H-bridge. The SMES will charge until 

the current flow through the inductor reaches steady-state. Then the SMES will switch to the steady-state 

mode of operation until backup power is required. Figure 182 shows the system used to simulate the SMES. 

Table 35 lists the settings in the model. Figure 183 shows the charging of SMES. The figure shows that the 

SMES steady-state current is about 10 kA and it takes around 400 s to reach steady-state.

After the SMES is charged, if a failure occurs in the system and backup power is needed, the SMES 

will switch to the discharge state. Figure 184 shows voltage across the load after a fault at 5 s. At 5.2 s, 

the SMES becomes the source for the load. Figure 185 shows the current flow through the load. 

Figure 184 shows that the SMES can create an over-voltage. The H-bridge can also be used to 

regulate the output voltage of the SMES. This is accomplished by rapidly switching between the 

discharge and steady-state modes of operation. A controller senses the output voltage of the SMES. If the 

voltage is reaching an unacceptable level, then the controller will switch the SMES to steady-state. The 

controller will monitor the voltage, and if it drops below the target voltage (within a set tolerance), then 

the controller will switch the SMES back to the discharge mode. The fast switching can create noise in 
NASA/CR—2015-218440 160

the output of the SMES. The amount of noise can be mitigated by increasing the output capacitance of the 

SMES. A small inductor can also be added to the output of the SMES to smooth current flow. Figure 186

and Figure 187 show the load voltage and current, respectively if voltage control is implemented.

TABLE 35.—SMES MODEL PARAMETERS 

Parameter Setting

Source voltage .................................................. 1000 V

Source resistance .................................................... 1 Ω

Source inductance............................................ 0.002 H

Load resistance .................................................. 8.33 Ω

SMES inductance ................................................. 64 H

SMES capacitance......................................1.3×10–3 °F

Figure 181.—SMES circuit and states.

Figure 182.—SMES Model.
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Figure 183.—SMES Voltage during Charging.

Figure 184.—SMES Voltage during Discharge.

Figure 185.—SMES Current during Discharge.

7.5 Power Converter Models

The types of power converters needed for the TePD system are inverters and rectifiers. Two 

topologies were considered for each type of power converter—a current source topology and voltage 

source topology.
NASA/CR—2015-218440 162

Figure 186.—SMES Voltage during Discharge with Voltage Control.

Figure 187.—SMES Current during Discharge with Voltage Control.

7.5.1 Rectifiers

Rectifiers are required in the system to covert AC power from the generators to DC power for the 

buses. 

7.5.1.1 Current Source Rectifier

The first topology examined was a modular current source converter, which is shown in Figure 188. 

The current source rectifier is designed for overvoltage protection. Also, the modular design allows 

modules to be added or deleted based upon the power requirements of the rectifier. However, this rectifier 

posed a number of challenges. First of all, limited information about the control of this rectifier is 

available. The control scheme is complex due to the number of switches and modules. Each module has 

to be individually controlled such that the output of the rectifier achieves the desired voltage. To simplify 

the process the module design shown in Figure 189 was used for the analysis (Ref. 184).

The control scheme used in the model was adapted from Solas (Ref. 184). The control scheme is 

PWM based. The PWM control consists of a control signal and triangular signal. A triangular signal is 

generated for each module. For each arm of the rectifier, four triangular signals are generated. 
NASA/CR—2015-218440 163

Figure 188.—Current Source Rectifier Topology.

Figure 189.—Current Source Rectifier Module.
NASA/CR—2015-218440 164

The phase of each wave is 90° apart. The phase shift between signals for each arm is 120°.

The reference signal is generated using the algorithm shown in Figure 190. The reference signal 

attempts to drive the output of the rectifier to the reference signal specified. 

Once the triangular and reference signals are generated, the algorithm shown in Figure 191 is used to 

determine the switching states of each module. A switching state of 1 for the upper modules means that 

the lower IGBT in the module is conducting and the upper is closed. A switching state of 0 for the upper 

modules means that the upper IGBT is conducting and the lower IGBT is closed. The opposite switching 

state occurs for the lower modules (i.e., a switching state of 1 means that the upper IGBT is conducting 

and the lower is not). 

The model used to test the rectifier designs is shown in Figure 192. The model was run using the 

parameters listed in Table 36. The output current and voltage are shown in Figure 193 and Figure 194. 

TABLE 36.—CURRENT SOURCE RECTIFIER MODEL PARAMETERS 

Parameter Setting

Source current...........................................................................75000 A

Source frequency ........................................................................ 400 Hz

Source resistance...............................................................................1 Ω

Source inductance ......................................................................0.001 H

Rectifier inductance (per inductor) ............................................0.017 H

Rectifier capacitance (per module) ...........................................0.0034 F

Rectifier output capacitance......................................................... 0.34 F

Switching frequency ............................................................... 18000 Hz

Kp ......................................................................................................0.5

Kp’........................................................................................................ 1 

Reference voltage........................................................................4500 V

Load resistance ............................................................................0.32 Ω

Figure 190.—Reference Signal Calculation.
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Figure 191.—Current Source Rectifier Switching Algorithm.

Figure 192.—Current Source Rectifier Model.
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Figure 193.—Current Source Rectifier Output Current.

Figure 194.—Current Source Rectifier Output Voltage.

A second problem with this type of rectifier is a high amount of losses. The input current and voltage 

(generated by a controlled current source) are shown in Figure 195 and Figure 196. Figure 195 

demonstrates that the input voltage must be extremely high in order to meet the target output voltage of 

the rectifier. The high number of switches in this topology leads to a high amount of switching losses. 

7.5.1.2 Voltage Source Rectifier

Due to the control and efficiency problems created by the current source rectifier, a voltage oriented 

controlled, voltage-source rectifier was considered. The voltage source topology is shown in Figure 197 

(Ref. 185).
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Figure 195.—Current Source Rectifier Input Current.

Figure 196.—Current Source Rectifier Input Voltage.

Figure 197.—Voltage Source Rectifier Topology.
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Figure 198.—VOC Control.

The control scheme for the voltage source rectifier is much less complex than the current source 

rectifier. The basic control scheme, called voltage oriented control (VOC), is shown in Figure 198 

(Ref. 186). The controller contains an outer PI voltage control loop with an inner current control loop. 

The voltage control attempts to regulate the output voltage of the rectifier to the specified value. The 

current loop strives to drive the current to a state of active power and minimizes the reactive component. 

The control scheme of this converter can help control bus voltage in a failure scenario. Also, the reduced 

number of switches greatly reduces the switching losses in the converter. However, scaling this converter 

is more difficult than the current source modular design. 

The VOC scheme selected for this rectifier uses the stationary dq reference frame. In order to use this 

reference frame, a coordinate transformation is required. The equations used to transform the coordinates 

are (Ref. 187):











































 −−

=



 





β

α

c

b

a

3

1

3

1 0

3

1

3

1

3

2





 





β

α

































−−

− =





















2

3

2

1

2

3

2

1

01

c

b

a





 





β

α





 





θθ−

θθ =



 





sin cos

cos sin

q

d





 









 





θθ

θ−θ =



 





β

α

q

d

sin cos

cos sin
NASA/CR—2015-218440 169

Where θ is the voltage angle which is found using a phase locked loop (PLL). A PLL is a feedback 

controller which “locks” two waveforms to the same frequency. This controller also has the ability to 

determine the frequency of a wave and find the phase between waveforms.

Once the coordinate transformation has taken place, the voltage and current signals are sent to the 

decoupled controller. Figure 177 shows the decoupled controller. As shown in the diagram, three PI 

controllers are required, two of which are current controllers. The proportional (kip) and integral (kii) 

coefficients of the current PI controllers are:

R F kL F k s ii s ip 10

2

; 10

2 π

<

π

<

Where Fs is the PWM switching frequency. The proportional (kvp) and integral (kvi) coefficients for the PI 

voltage controller are:

; 001.0

30

2 = π

< vi

m

s vp k

E

CF k

The integral portion of the PI voltage controller has almost no effect on the outcome of the model. 

This is why a small coefficient is arbitrarily chosen. The proportional coefficient for the voltage controller 

has a very strong influence on the performance of the controller. If the value of the coefficient is too 

large, the model will not be able to converge on a solution or will not be able to reach the target voltage. 

If the value selected is too small, a large amount of harmonic distortion will be present in the bus voltage. 

In some cases, there may be a large step in the reference voltage. This large step will cause the 

controller to demand a higher voltage than the rectifier can supply; therefore, a saturation block needs be 

added to the current controller to ensure that the reference voltage does not exceed the maximum voltage 

output of the rectifier. This issue can also arise for the voltage controller, so a saturation block will also be 

used in conjunction with it. Although using saturation fixes the problem of demanding too large of a 

voltage or current, it introduces another problem. When the voltage or current is limited, a phenomenon 

called integrator wind-up can occur. This causes an overshoot in the response of the PI controller and the 

controller error will increase. In order to correct this problem, the error input into the current controller 

should become 

k p

− vv

+ε=ε

where ε is the limited error and v is the limited voltage. The same principle can be applied to the 

voltage controller.

The voltage source rectifier was tested using the model shown in Figure 199 and the parameters listed 

in Table 37. The input current and voltage of the rectifier are shown in Figure 200 and Figure 201. The 

output current and voltage is shown in Figure 202 and Figure 203. The output does contain some 

harmonic interference. This response can be smoothed by fine tuning the controller parameters or 

increasing the output capacitance. 
NASA/CR—2015-218440 170

TABLE 37.—VOLTAGE SOURCE RECTIFIER MODEL PARAMETERS 

Parameter Setting

Source voltage ...........................................................................70000 V

Source frequency ...........................................................................60 Hz

Source resistance .............................................................................. 1 Ω

Source inductance.......................................................................0.001 H

Rectifier inductance ..................................................................0.0006 H

Rectifier output capacitance.......................................................... 0.34 F

Switching frequency ................................................................18000 Hz

αv (PI control variable).................................................................. 10000

αi (PI control variable).................................................................100000

Reference voltage ........................................................................4500 V

Load resistance ............................................................................ 0.32 Ω

Figure 199.—Voltage Source Rectifier Model.

Figure 200.—Voltage Source Rectifier Input Current.
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Figure 201.—Voltage Source Rectifier Input Voltage.

Figure 202.—Voltage Source Rectifier Output Current.

Figure 203.—Voltage Source Rectifier Output Voltage.
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The voltage source rectifier was chosen for the system studies since the controller has proved to be 

more stable than that of the current source rectifier. Furthermore, the losses are much smaller with the 

voltage source design.

7.5.2 Inverters

As in the case of the rectifier, two inverter topologies were considered – current source and voltage 

source. 

7.5.2.1 Current Source Inverter

The topology shown in Figure 204 is the current source inverter. This converter is bi-directional, 

meaning that it can also function as a rectifier. The pulses to control the switches are created using a 

PWM generator. The PWM signals are created by comparing a carrier signal to a control signal. The 

carrier signal is a triangle wave that is set to a high frequency. In this example, it was set to 20,000 Hz. 

The control signal is a sinusoidal signal at the desired output frequency. These signals are shown in 

Figure 205. The blue signal is the carrier signal and the green signal is the control signal. The magnitude 

of the carrier signal corresponds to the modulation index. The modulation index determines the amplitude 

of the output voltage of the inverter. The amplitude of the control signal should not be greater than that of 

the carrier signal to ensure that overmodulation does not occur. The pulses sent to the switches are created 

by comparing the carrier signal to the control signal. Based upon this comparison, the signal is set to 0 or 

1. For each arm of the bridge, two signals are sent. These signals should be opposite of each other. An 

example of the pulses is shown in Figure 206. A set of pulses is generated for each of the three arms of 

the inverter.

Figure 204.—Current Source Inverter Topology.
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Figure 205.—PWM Control and Carrier Signals.

Figure 206.—PWM switching pulses. 

TABLE 38.—CURRENT SOURCE INVERTER MODEL PARAMETERS

Parameter Setting

Source current (DC).....................................................................1200 A

Switching frequency ................................................................ 10000 Hz

Modulation..........................................................................................0.8

Output frequency.......................................................................... 400 Hz

Inverter capacitance ...................................................................6×10–4 F 

Inverter inductance.................................................................... 0.0012 H

Load power ........................................................................... 5400000 W

The model was run using the parameters shown in Table 38. The input voltage to the inverter is shown in 

Figure 207. The output voltage and current of the inverter are shown in Figure 208 and Figure 209. 
NASA/CR—2015-218440 174

Figure 207.—Current Source Inverter Input Voltage.

Figure 208.—Current Source Inverter Output Voltage.

Figure 209.—Current source inverter output current. 
NASA/CR—2015-218440 175

7.5.2.2 Voltage Source Inverter

Figure 208 and Figure 209 show that the output of the current source inverter is noisy. For this 

reason, a voltage source inverter topology was considered. The circuit of voltage source inverter with a 

load is shown in Figure 210. The model is shown in Figure 211. The inverter model consists of a 

capacitor connected across the terminals of a three phase IGBT/diode bridge. Like the current source 

inverter, the voltage source inverter is PWM controlled. The model was run using the parameters listed in 

Table 39. The input current for the inverter is shown in Figure 212. (The current can be smoothed by 

using an inductor in series with the voltage source.) The output current and voltage of the inverter is 

shown in Figure 213 and Figure 214. The response of this inverter topology is much smoother than that of 

the current source inverter. For this reason, the voltage source topology was selected for the system 

modeling. 

TABLE 39.—VOLTAGE SOURCE 

INVERTER MODEL PARAMETERS

Parameter Setting

Source voltage, V................................................................12000 

Switching frequency, Hz.....................................................10000 

Modulation...............................................................................0.8

Output frequency, Hz..............................................................400

Load power, W ...............................................................5400000

Figure 210.—Voltage Source Inverter Topology.

Figure 211.—Voltage Source Inverter Model.
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Figure 212.—Voltage Source Inverter Input Current.

Figure 213.—Voltage Source Inverter Output Current.

Figure 214.—Voltage Source Inverter Output Voltage.
NASA/CR—2015-218440 177

7.6 System Modeling

The components described in the previous sections were used to construct the system models. A 

variety of system models were created to simulate several failure conditions. The models were created in 

Simulink using SimPowerSystems. (The state-space models were not used because some of the dynamic 

behavior of the system would be lost.) All the models are run using a MATLAB script, and all variables 

in the model can be changed by the script. 

The generator in the models was represented by a voltage source with an inductance and resistance in 

series. The generator parameters used for the system models are shown in Table 40. 

The motors were modeled using SimPowerSystem’s permanent magnet machine model. The motor 

model can be speed or torque controlled. The parameters that were set in the model are shown in Table 41. 

All component variables have the settings that were listed in the component model summaries.

The naming convention used for all components in the Simulink models and MATLAB scripts are 

shown in Figure 215. The figure also shows which SSCB are open and closed during normal operation. 

The same convention will be used the failure scenario diagrams. In the failure models, the SSCB will 

switch from the normal operation state to the state shown in the failure diagram after a set delay from the 

time of the fault. The breakers that switch and the switching times are dictated by the MATLAB script.

TABLE 40.—SYSTEM MODEL GENERATOR PARAMETERS

Parameter Setting

Voltage amplitude, V ..........................................................25000

Frequency, Hz...........................................................................60

Resistance, Ω.......................................................................0.001

Inductance, H .........................................................................0.01

TABLE 41.—SYSTEM MODEL MOTOR PARAMETERS (REFS. 188 AND 189) 

Parameter Setting

Speed, rpm................................................................................................................................................ 3000

Armature inductance, H ..................................................................................................................... 0.000102

Armature resistance, Ω................................................................................................................................ 0.2

Pole pairs........................................................................................................................................................ 4 

Amplitude of the flux induced by the permanent magnets of the rotor in the stator phases .............. 0.044235

Combined viscous friction of rotor and load, N.m.s.............................................................................. 0.1035

Combined inertia of rotor and load, kg/m2 .......................................................................................... 0.00112
NASA/CR—2015-218440 178

Figure 215.—System Naming Convention.

7.6.1 Single Motor Model

The single motor model represents the interactions of the components along a single path from a 

generator to one motor. Faults can be placed in the path and the response of the system can be tracked. To 

demonstrate the model capabilities, the model was run with a line to ground fault on the bus that occurs at 

0.6 s. Figure 216 and Figure 217 show that the source is mostly unaffected by the fault. Figure 218 

shows the fall in bus voltage after the fault. It takes the bus about 0.04 s to drop to zero. Figure 219 and 

Figure 220 show the bus current upstream and downstream of the fault. The figures demonstrate the 

magnitude of the fault current which is about 5 times the nominal current. Figure 220 shows a delay 

between the circuit breakers isolating the fault and the time that the SMES activates. The SMES current is 

not filtered in this case. Filtering can be added to smooth the bus current when the SMES is discharging. 

Figure 221 and Figure 222 show the load voltage and current. The delay between the fault and the SMES 

discharge is again present. In this simulation, voltage regulation for the SMES was not active; therefore, 

an overvoltage situation occurs.
NASA/CR—2015-218440 179

Figure 216.—One Motor Simulation Source Current (A).

Figure 217.—One Motor Simulation Source Voltage (V).
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Figure 218.—One Motor Simulation Bus Voltage (V).

Figure 219.—One Motor Simulation Bus Current Upstream of the Fault (A).
NASA/CR—2015-218440 181

Figure 220.—One Motor Simulation Bus Current Downstream of Fault (A).

Figure 221.—One Motor Simulation Load Voltage (V).
NASA/CR—2015-218440 182

Figure 222.—One Motor Simulation Load Current (A).

7.6.2 Fault Isolation Model

The fault isolation model simulates the response of one branch of the system. A branch consists of a 

generator, rectifier, bus, four inverters, four motors, and any required protection devices. The model can 

be used to track the response of the branch if a fault occurs. This model will only allow the effect of a 

fault to be examined; it does not include recovery. This model serves as the base to the models created to 

simulate a variety of failure scenarios.

7.6.3 Nominal Recovery Model

The nominal recovery model is the fault isolation model with an emulation of the L-1 branch and 

right side of the architecture. The generator, rectifier, and bus for the L-1 branch are emulated using a DC 

voltage source with an added inductance. The L-1 branch loads were modeled as resistors. Resistive loads 

were chosen since the power factor of the superconducting motors should be close to unity; however, an 

inductance can be added to the model. Simply change the load block to a RL load. The inductance will 

automatically be pulled from the MATLAB script. 

The emulation of the right side of the architecture is dependent of the failure scenario being modeled. 

In the case of a failure on the right side of the architecture, the right side will be emulated as a load. In all 

other scenarios, the right side of the architecture will be emulated as a voltage source. The model can be 

viewed in Appendix B. The model is used as the base to simulate the failure scenarios.

7.6.4 Failure Scenarios

7.6.4.1 Right Engine Failure Model

The right engine model simulates the system response in the event that the right engine fails. The 

model consists of the nominal recovery model with a load that represents the right half of the electric grid. 

An overview of the failure scenario is shown in Figure 223. When the right engine fails, power from the 

left engine must be rerouted to supply power to the right engine loads. L-1 is used to power its loads and 

the loads of L-2. The power from L-2 is routed to the R-1 and R-2 loads. Energy storage is used to power 

R-1 and R-2 for the time that it takes to reroute the power from GL-2. 
NASA/CR—2015-218440 183

The right engine failure model was run with a failure occurring at 0.025 s. The source and bus voltage 

on the L-2 branch is similar to that of the single motor model. The biggest difference is now the power is 

rerouted to the right side of the architecture during the failure. The current flow to the right side load 

emulation is shown in Figure 224. The current is shown as zero before failure because the right side 

source is not simulated in this case. After power is rerouted, the load current reaches steady state after 

about 0.005 s. When the failure occurs, the SMES is discharged to supply power to the loads while power 

is being rerouted. The SMES discharge current is shown in Figure 225. Since the power is rerouted from 

L-2 to the right side of the architecture, L-1 must now supply the L-2 loads. The input current to the 

inverters to the L-2 loads is shown in Figure 226. It appears that the emulated source for L-1 has difficulty 

supplying the current to the L-2 loads in this situation. The unsteady and low amount of current causes the 

motors to lack input power. The input voltage and current for the motors are shown in Figure 227 and 

Figure 228. Further study is needed to determine how to emulate the L-1 load so that the correct level of 

power is delivered to the L-2 loads in the case of the right engine failure. Also, the shared power from L-1 

causes a drop in current supplied to the L-1 loads, which is shown in Figure 229. 

Figure 223.—Right Engine Inoperable Power Flow.

GL-2

SCFCL

Energy 

Storage

SCFCL

SCFCL

N.O.

ML-3 MR-3 ML-6 MR-6

Left 

Engine

Load/

Source

Emulation

L-1 Branch 

Emulation

Load 

Emulation

Load 

Emulation

Load 

Emulation

Load 

Emulation
NASA/CR—2015-218440 184

Figure 224.—Right Engine Failure—Right Engine Load Emulation Current (A).

Figure 225.—Right Engine Failure SMES Discharge.
NASA/CR—2015-218440 185

Figure 226.—Right Engine Failure L-2 Inverter Input Current (A).

Figure 227.—Right Engine Failure Motor Input Voltage (V).
NASA/CR—2015-218440 186

Figure 228.—Right engine failure motor input current (A).

Figure 229.—Right Engine Failure L-1 Load Current (A).
NASA/CR—2015-218440 187

7.6.4.2 Left Engine Failure Model

The left engine failure model is the same as the right engine failure model except the right side of the 

grid is now modeled as a source rather than a load. A diagram of the left engine out scenario is shown in 

Figure 230. Like the L-1 emulation, the right side of the grid is modeled as a DC voltage source and 

inductance. When the left engine fails, the L-1 and L-2 loads must be powered by the right side of the 

grid. The SMES is used to power the loads for the short amount of time that it takes to reroute power 

from the right side of the grid to the L-1 and L-2 loads. 

The engine out condition was simulated using this model. When the engine fails, the source current 

and voltage are driven to zero. The results of the simulation are shown in Figure 231 to Figure 235. In this 

simulation, the engine failure occurs at 0.089 s. Figure 231 shows that the bus voltage rapidly falls after 

the failure; however, there is a rapid increase in voltage once power is rerouted. Figure 232 shows the bus 

current. After the failure, bus current falls to zero after about 0.0005 s. The SMES discharge current is 

shown in Figure 233. Figure 234 shows the inverter input current. Current flow is actually reversed after 

the failure. This may be due to a back EMF being produced by the motor. More study is needed to 

determine the exact cause of this phenomenon. Figure 235 and Figure 236 show the motor current and 

voltage. The motors lose a great deal of power during the failure. More analysis is needed to determine 

the cause of the power loss and how to update the recovery scheme in order to return the motors to normal 

operation. 

Figure 230.—Left Engine Inoperable Power Flow.

GL-2

SCFCL

Energy 

Storage

SCFCL

SCFCL

N.O.

ML-3 MR-3 ML-6 MR-6

Left 

Engine

Load/

Source 

Emulation

L-1 Branch 

Emulation

Load 

Emulation

Load 

Emulation

Load 

Emulation

Load 

Emulation
NASA/CR—2015-218440 188

Figure 231.—Left Engine Bus Voltage.

Figure 232.—Left Engine Failure Bus Current.
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Figure 233.—SMES Discharge Current.

Figure 234.—Left Engine Failure Inverter Input Current.
NASA/CR—2015-218440 190

Figure 235.—Left Engine Failure Motor Input Current.

Figure 236.—Left Engine Failure Motor Input Voltage.
NASA/CR—2015-218440 191

7.6.4.3 L-2 Branch Fault Model

The L-2 branch fault model is the nominal recovery model with a fault inserted on the L-2 bus. (The 

fault block can be moved to other locations on L-2 to simulate other scenarios.) A diagram explaining this 

scenario is shown in Figure 237. When the fault occurs, power is rerouted from the L-1 branch to the L-2 

loads. A demonstration of the fault model is shown in Figure 238 to Figure 243. In this case, the system 

was faulted at 0.04 s. Figure 238 demonstrates the fall of bus voltage after the occurrence of the fault. 

Figure 239 shows the increase in current on the bus upstream of the fault. Figure 240 shows the fault 

current on the bus. The fault current is about 3 times the level of the nominal current. Figure 241 shows 

the input current into the inverter. A delay between the time of the fault and when the SMES begins to 

supply current to the load is shown in the figure. Again, the SMES current can be smoothed with the use 

of inductive filters. Figure 242 and Figure 243 show the load voltage and current. Again the time delay 

between the fault and recovery is present. Also, with the component parameters settings, the input power 

for the load is reduced during recovery. 

Figure 237.—L2 Branch Fault Power Flow.

GL-2

SCFCL

Energy 

Storage

SCFCL

SCFCL

N.O.

ML-3 MR-3 ML-6 MR-6

Left 

Engine

Load/

Source 

Emulation

L-1 Branch 

Emulation

Load 

Emulation

Load 

Emulation

Load 

Emulation

Load 

Emulation
NASA/CR—2015-218440 192

Figure 238.—L-2 Branch Fault Simulation Bus Voltage (V).

Figure 239.—L-2 Branch Fault Simulation Bus Current Upstream of Fault (A).
NASA/CR—2015-218440 193

Figure 240.—L-2 Branch Fault Simulation Bus Current Downstream of Fault (A).

Figure 241.—L-2 Branch Fault Simulation Inverter Input Current (A).
NASA/CR—2015-218440 194

Figure 242.—L-2 Branch Fault Motor Input Voltage (V).

Figure 243.—L-2 Branch Fault Simulation Motor Input Current (A).

7.6.4.4 L-1 Branch Fault Model

Like the L-2 branch fault model, the L-1 branch fault model is the nominal recovery model with a 

fault (Figure 244). This time the fault occurs on the L-1 branch. When the fault occurs, power is routed 

from the L-2 branch to the L-1 loads. Energy storage is used to power the L-1 loads for a short amount of 

time. The behavior of this fault scenario will mimic the behavior of the L-2 branch during a fault. 
NASA/CR—2015-218440 195

Figure 244.—L1 Branch Fault Power Flow.

7.6.5 Running the Models

The first step in running any of the models is to first run the corresponding m-file. The models will 

not run properly unless the correct corresponding script is used. Once the script runs, the Simulink model 

can be run. Scopes can be added to the model to track the system states. However, the addition of scopes 

will slow the model.

The full system dynamic simulations are computationally intensive and require a large amount of 

computer speed and run-time. Using a 64-bit machine with a minimum of 4.00 GB of ram is 

recommended for running the models. With a 64-bit machine with 8.00 GB of ram and a 3.4 GHz 

processor, a simulation of 0.1 s using the recovery model takes approximately 4 days. The single motor 

model can be run in approximately 3 hr. 

One way to reduce the amount of time needed for the simulations is to start the models at steady-state. 

One way to achieve this is to run the nominal recovery model to steady-state and save the simulation 

state. To save the simulation state, go to the configuration parameter window and navigate to the data 

import/export pane. Select the final states check box, and then select the save complete SimState in final 

state check box. Enter a variable name for the SimState; then run the model long enough for the system to 

reach steady state. Once these results are saved, the simulation can start from this point. To resume the 

GL-2

SCFCL

Energy 

Storage

SCFCL

SCFCL

N.O.

ML-3 MR-3 ML-6 MR-6

Left 

Engine

Load/

Source 

Emulation

L-1 Branch 

Emulation

Load 

Emulation

Load 

Emulation

Load 

Emulation

Load 

Emulation
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simulation from the saved point, again go to the configuration parameters window. In the data 

import/export pane, select the initial state box under load from workspace. Enter the name of the variable 

used to save steady-state simulation. Keep the start value at the same start time. For the stop time, use the 

original stop time plus any additional time needed to simulate the failure. The different failure scenarios 

can then be studied by immediately causing the fault or failure. 

7.7 Summary and Future Studies

During this study, a variety of component and system models were developed. The models will help 

determine component and system responses under steady-state and failure scenarios. Some important 

parameters that can be extracted from the simulations are fault currents and recovery time. Also, 

protection components can be activated and deactivated in order to determine the necessary level of 

protection in the system. 

While the models are a good representation of the system, a number of improvements can be made to 

increase the fidelity of the simulations. One improvement needed for the SFCL model is to further 

examine the temperature calculation. This will include determining whether to use the thermal equivalent 

circuit approach or to use heat transfer equations. Also, the current model assumes that the amount of 

coolant is large enough that its temperature will remain at 77 K. This may not be the case and the model 

will need to be altered. Also, a copper shunt can be included in the model that was presented. Often 

resistive SFCL use a copper shunt in parallel. The shunt helps smooth the temperature rise in the 

superconductor during quenching to reduce the risk of damage from the heat being generated. 

For the SSCB model, the inverse-time overcurrent protection scheme can be improved. In many 

applications, a rotating disc model is used in the control. In this case, the timer is not completely reset 

when the current falls below the tripping current. Instead, a “disc” with some inertia is slowed to a stop. 

The disk will not immediately stop, so if an over-current is detected again, the disk will already have 

some speed and ramp up to the tripping condition faster (Ref. 190). Lastly, the possibility of constructing 

a state-space model of the SSCB will be investigated.

The SMES model could be refined by creating a more sophisticated overvoltage protection scheme. 

The current scheme can create noise in the output response. The use of a surge arrester in the circuit could 

be studied as another means of protection. 

The current source rectifiers and inverters still need to be studied to find a stable control scheme that 

minimizes the losses in the component. The control of advanced power electronics is an ongoing area of 

research. New approaches to the control of these types of converters will likely emerge in the near future. 

Updated control schemes can be incorporated into the base model to further test the current source 

converter topologies.

The most significant improvements to this system model can be achieved by creating a higher fidelity 

generator and motor models. This would entail detailed component modeling for the machines and then 

incorporating the new models into the existing system models. Fan maps can also be included in the 

model to better predict the speed or required torque for the motors.

Another area of improvement for the system model is increasing model speed. One possibility for 

decreasing simulation time is to use state-space models. However, the state-space models for the 

converters would average the response of the converter over several switching periods, and some of the 

dynamic response would be lost. Due to this problem, further study is needed to determine whether using 

the state-space models would accurately portray the response of the system.
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8.0 Conclusions

Electrical system integration requirements are immature for revolutionary electric propulsion systems. 

As such, TeDP component technology development requires assumptions to be made regarding the 

voltage levels, regulation, and protection requirements when assessing concept performance. Therefore, 

this study represents an effort to provide initial justification for the selection of TeDP architecture voltage 

levels and limits. The system of interest for this study was the N3-X superconducting DC architecture. 

However, the models and tools developed in this study provide many of the building blocks for 

addressing voltage selection for variety of electric propulsion system architectures with varying 

requirements and compositions. As these tools are applied to maturing TeDP concepts, it is hoped that the 

methods applied in this study will begin to frame the development of voltage standards for airborne 

propulsive power systems.

The voltage envelopes are consistent throughout the electrical standards literature review. At the point

of interface, the voltage can drop to zero for a period of time followed by a recover time and recovery limit. 

The boundaries of the transient and abnormal voltage regulation will be determined from the regulation and 

protection system’s capabilities. Improvement in voltage regulation will come at the expense of adding 

filtering thereby increasing weight. The voltage quality may also be driven by requirements from operators 

and regulatory bodies to control issues such as EMI. The literature review on voltage standards also brought 

attention to specifying voltage and frequency together. This can be prudent as an overvoltage condition at 

the same time as an underfrequency condition can result in overfluxing of the electric machines which will 

lead to damaging eddy currents and excessive heating in the machine core. The voltage standards for 

isolated microgrids and for grids with a high penetration of distributed sources also defined the power factor 

of the loads in establishing voltage response during a transient. As load characteristics are better defined, the 

converter controls and filtering will be revisited to ensure that the converters meet the required voltage 

regulation. The IEEE recommended practice for MVDC power systems on ships provides several area of 

further work that will be necessary to create a voltage standard. Areas include the categorization of loads, 

limits of fault conditions on the equipment, stability studies and safety cases.

Parametric sizing models for all components within NASA’s N3-X TeDP electrical system were 

generated and exercised to determine the optimal operating voltage for the system. Mass and efficiency 

sensitivity was evaluated for superconducting generators, AC and DC cable runs, power conversion 

equipment, superconducting fault current limiter, solid state circuit breakers, and a superconducting 

magnetic energy storage system. The combined sensitivity for component mass and efficiency were 

evaluated assuming general cryocooling mass sensitivity assumptions. For the baseline bi-polar DC 

architecture configuration, in all cases the optimal operating voltage was found to be less than ±4.5 kV. 

Systems which rely on converters for isolation and redundancy for recovery, as opposed to breakers and 

energy storage, the optimal voltage was found to be ±2 kV. Additionally, the system mass is insensitive to 

voltage on a fairly reasonable voltage range. 

Observations and outstanding questions related to the TeDP system components modeled in this 

study provide potential areas of further consideration in the development of the electrical systems 

architecture.

8.1 Power Electronics

The preferred voltage ranges selected for this architecture are unique to the architecture selected and 

the component performance assumptions made. Voltage sensitivity analysis showed that the mass and 

efficiency of the semiconducting drive the voltage sensitivity of the overall systems. Considering mass 

penalties for cryocooling, the efficiency trends for IGBTs and diodes have the largest impact on mass 
NASA/CR—2015-218440 198

sensitivity. As such, for an airborne high power DC TeDP system, light weight and highly efficient power 

conversion equipment is a pacing technology. 

Current Source Converters (CSCs) were chosen to rectify the AC generator voltage and invert the DC 

power to the propulsor motors. Current source converters use inductors to store energy and that 

architecture inherently limits the current giving the system an additional level of protection in the event of 

a fault. The source converters were unidirectional and the load converters where bi-directional. Both 

converter topologies showed the ideal voltage in the range between ±3 to ±6.5 kV. IGBTs do not have the 

ability to block higher voltages so as the voltage increased more IGBTs were needed to effectively block 

the voltage, increasing the weight. 

At low voltages, the current in the system required multiple IGBTs to be in parallel driving up the 

weight. Some of the IGBT and diode parameters used to model the weight and efficiency sensitivities were 

derived from extrapolating normally conducting devices down to superconducting temperatures so using 

modelling data near the freeze out temperature would improve the modelling. The freeze out temperature is 

the temperature low temperature beyond which semiconductors will not function and is typically determined 

by the ionization energy of the dopants as well as the doping concentration of the semiconductors

(Ref. 191). Heavy doping may allow devices to operate below the freeze out temperature and allow 

semiconducting devices to operate in the same environment as the rest of the TeDP power system. Further 

characterization of semiconducting materials and types over a range of temperature will provide more model 

fidelity. This additional characterization would also work to reducing the switching losses as the energy turn 

on and turn off is directly related to the energy losses of the IGBTs. Models that use first principles to model 

the behavior of IGBTs and diodes at low temperature will improve the weight and efficiency estimates of 

the converters. Finally, the converter models will need to incorporate filtering elements to reduce Electronic 

Magnetic Interference (EMI) to a reasonable level and maintain power quality. Filtering can add significant 

weight to the converters but can also be used to improve system stability. 

8.2 Protection

The protection of the superconducting TeDP electrical system will be a challenge due to speed that 

faults and disturbances occur while maintaining power to the propulsors. The speed that a circuit breaker 

can interrupt a fault is directly related to the amount of energy that requires dissipation during a 

disturbance. The slower the fault interruption, the more energy has to be absorbed by the cryogenic 

system. In turn, leading to increases in the size of the cryo-system. Also, fast circuit breakers may reduce 

the number of required fault current limiters or eliminate them altogether. Solid-state circuit breakers 

(SSCBs) were chosen for this study due to the fast operating time and ability to operate at near cryogenic 

temperatures. The drawback to SSCBs is that they have higher conduction resistances than 

electromagnetic circuit breakers or hybrid circuit breakers. The higher losses of SSCB may be sufficient 

to perform a trade in the efficiency of the system during normal operation and the efficiency penalty of 

carrying extra weight from a larger cryo-system and SFCLs if slower circuit breakers are used. The dual 

use of the converters as regulation and protection may eliminate both the circuit breakers and fault current 

limiters though some isolation equipment may still need to be necessary to physically separate electric 

machines and energy storage from the network for maintenance or after reconfiguration. 

8.3 Energy Storage 

The dynamic models are also crucial in identifying the requirements on energy storage devices. For 

this architecture, the role of energy storage in this TeDP architecture was limited to that of an 

uninterrupted power system to support the loads during a source fault. In order to eliminate interfaces 

between room temperature and cryogenic electrical systems, superconducting magnetic energy storage 
NASA/CR—2015-218440 199

was selected for UPS power. The SMES benefits from the ability to rapidly discharge the stored energy 

and is shown to exhibit fairly high energy densities when assuming the use of high strength structural 

materials. This energy density increases with the amount of energy stored. Additional work is necessary 

to develop discharge management and control logic so the SMES can provide DC power to the loads 

during a fault in a regulated manner. 

The SMES is attractive as a high energy and power dense system which operates at cryogenic 

temperatures. However, it also suffers from several deficiencies. First, the weight of the power electronics to 

allow for charging and discharging of the SMES introduce detriments to the overall system mass. Second, 

SMES cryocooling requirements also increase the overall system mass. Third, the volume of a SMES device 

may be limited for a high energy airborne application. Fourth, the mass and length of the superconductor for 

high energy, high voltage, SMES devices may introduce prohibitive costs and manufacturing challenges. 

Lastly, concepts for fault tolerant SMES designs are needed which can dissipate the massive amounts of 

energy within the inductive coil in a controlled fashion when a fault occurs. 

8.4 Distribution

Cables systems are practically negligible in contributions to mass and efficiency for this DC system. 

Additionally, as with all the other systems, the insulation and cryogenic cooling systems contribute more 

to the overall mass than the superconductor and dielectric insulation. As such, for both AC and DC 

superconductor, the ratio of operating and fault currents to the critical current can be selected so as to 

minimize the losses at a very small detriment to cable weight. The N3-X architecture concept was 

prescribed with a DC distribution system. As such, the losses and masses associated with power 

conversion were adopted in the system. However, by operating at higher critical current ratios of the 

distribution cables, superconducting AC TeDP electrical system concepts may begin to look more 

promising. However, these concepts would require additional analysis of the implications of regulation, 

protection, and recovery. 

8.5 Dynamic Modeling

During the voltage sensitivity analysis assumptions were made regarding the overcurrent and 

regulation requirements for the system. Therefore, dynamic models of the entire system were made to 

define voltage regulation requirements and simulate critical fault scenarios to determine the isolation, 

protection, and recovery strategies for the system. 

The dynamic models developed in this project will be paramount in establishing the fault current and 

undervoltage that the equipment will see during disturbances and how fast those faults will propagate 

through the system. That will help future work in sizing the protection equipment and the speed at which 

protection equipment needs to act to isolate failures. Initially the dynamic models will help set the limits 

around current and voltage during faults so that the faulted lines are removed from operation and the 

models can be then used to tune the converter controls and protection settings so that the system does not 

disconnect for high motor start currents. Protection zones can be established once the dynamic models can 

give indication of speed of failures and the usability of power electronics in the protection scheme. The 

analysis of reducing the size and weight of the system using different converter topologies and control 

schemes to determine if it is viable to use the power electronics for protection will utilize dynamic models 

for failure and recovery scenarios. 

The dynamic models may also be used to study the impact that converter filtering and distribution 

line impedances have on stability. Literature review of DC systems showed the importance of 

understanding the system impedances especially around bi-directional converters. The fact that the 

current can flow in both directions increases the risks of instability and the possibility of using those 

converters for protection in both directions will requires careful study.

NASA/CR—2015-218440 201

Appendix A.—Acronyms and Abbreviations

AC Alternating current

BPS Bypass switch

DC Direct current

CSC Current-source converter 

EDR Engineering Department Report 

EMCB Electromechanical circuit breaker

EPACS Electrical Power and Control Systems

EPRI Electric Power Research Institute

ESR Equivalent series resistance 

ETL Energy Transfer Line

ETO Emitter turn-off thyristor

GE General Electric

GTO Gate turn-off thyristor

HCB Hybrid circuit breaker

I.D. Inner diameter

IEC International Electrotechnical Commission 

IEEE Institute of Electrical and Electronics Engineers

IGBT Insulated gate bipolar transistors

IGCT Integrated gate-commutated thyristor

Jc Critical current density

Je Engineering critical current density 

kV Kilovolt 

MGT MOS gated thyristor 

MCT MOS controlled thyristor

MOS Metal Oxide Semiconductor

MOV Metal oxide varistor

NERC North American Electric Reliability Corporation 

NPT Non-punch through 

O.D. Outer diameter

POI Point of interconnection 

PSS Power system stabilizer

PT Punch through 

PWM Pulse-width modulation 

RMS Root Mean Square 

RTAPS Research and Technology for Aerospace Propulsion Systems 

SCR Silicon Controller Rectifier

SEI Sumitomo Electric Industries 

SFCL Superconducting fault-current limiters

SMES Superconducting Magnetic Energy Storage 

SSCB Solid-state circuit breaker
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TeDP Turboelectric Distributed Propulsion 

TG Trench-gate

TRL Technology readiness level

UPS Uninterrupted power supply 

UTC University Technology Center

Vdc Volts direct current

VSC Voltage source converter
Appendix B.—IGBT Data

TABLE B.1.—ROOM TEMPERATURE IGBT DATA FOR EXISTING IGBTS. 

Manufacturer ABB ABB Infineon Mitsubishi Mitsubishi Mitsubishi Mitsubishi Mitsubishi Mitsubishi Mitsubishi Mit

IGBT Part Number 5SNA0600G

650100

5SNA1200G

450300

FZ600 

R65KE3

CM1200HA50HCM750HG130RCM900HG90HCM450DY24SCM800HB66HCM900HC90HCM1500HC66RCM1

Nominal current, A 600 1200 600 1200 750 900 450 800 900 1500 

Over current, A 1200 2400 1200 2400 1500 1800 900 1600 1800 3000 

Surge current, A 6000 9000

Over current withstand duration, ms 10 10 1

Nominal blocking voltage, kV 6.5 4.5 6.5 2.5 6.5 4.5 1.2 3.3 4.5 3.3 

Conduction voltage drop, V 3.5 3 3 3 5 3.3 2 4.2 3.5 3.2 

Current rating for conduction voltage 

drop, A

600 1200 600 1200 750 900 450 800 800 1500 

Gate drive resistance, Ω 3.9 1.5 1.3 2.5 3.9 10 8 2.5 10 1.6 

Time to turn off, μs 2.16 2.88 7.7 3.5 8 7.2 0.9 3.5 7.2 3 

Time to turn on, μs 0.91 0.96 1.03 3.6 1.23 3.6 1 3.6 3.6 1.28 

Turn off energy, J 1.95 4.96 2.9 0.86 3.1 2.5 0.048 1.2 2.5 2 

Turn on energy, J 3.8 3.08 3.4 1.5 3.35 4.2 0.0549 1.7 4.2 2.1 

Turn on di/dt, A/μs 2600 4800 2.9 1.2

Overcurrent withstand relative to 

nominal

2 2 2 2 2 2 2 2 2 2 

Conduction loss - DC/DC duty cycle, 

Wa

1050.00 1800.00 900.00 1800.00 1875.00 1485.00 450.00 1680.00 1771.88 2400.00 

Conduction loss - PWM for 

inverter/rectifier, Wb

1073.62 1840.50 920.25 1840.50 1917.18 1518.41 460.12 1717.80 1780.57 2453.99 

Switching loss - DC/DC converter, Wc 57500.00 80400.00 63000.00 23600.00 64500.00 67000.00 1029.00 29000.00 67000.00 41000.00 

Switching loss - PWM control, Wc 18302.82 25592.11 20053.52 7512.11 20530.99 21326.76 327.54 9230.99 21326.76 13050.71 

Semiconductor power loss - DC/DC 

converter, W

58550.00 82200.00 63900.00 25400.00 66375.00 68485.00 1479.00 30680.00 68771.88 43400.00 1

Semiconductor power loss - PWM 

control, W

19376.44 27432.61 20973.77 9352.61 22448.17 22845.17 787.66 10948.78 23107.33 15504.70 

Maximum collector dissipation, W 11900.00 10500.00 12500.00 10400.00 10400.00 11300.00 3330.00 10400.00 11900.00 15600.00 

Mass, kg 1.76 1.76 1.4 2.2 1.4 1.35 0.58 1.5 1.5 1.2 

Freewheeling diode ABB ABB Infineon Mitsubishi Mitsubishi Mitsubishi Mitsubishi Mitsubishi Mitsubishi Mitsubishi Mit

Part Number 5SNA0600G

650100

FZ600 

R65KE3

CM1200HA50HCM750HG130RCM900HG90HCM900HC90HCM1500HC66RCM1

Forward voltage, V 3.2 3 2.8 3.3 4 4.8 2.2 

Rated current for forward voltage, A 600 600 1200 750 900 900 1500 

Reverse recovery energy, J 1.1 1.1 1 1 1 1.05 

Approx. diode resistance, Ω 0.005333333 0.005 0.002333333 0.0044 0.004444444 0.005333333 0.001466667 

Data used to populate this table is from datasheets available from the manufacturer’s website (accessed Jan-March 2014). a Calculated for DC/DC Duty Cycle = 0.5

b Calculated for PWM duty = -0.64 (negative for rectifier operation) c Calculated for switching frequency = 10 kHz

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Appendix C.—Dynamic Models

An overview of the dynamic nominal recovery model is shown in this appendix. The right side of the 

architecture is emulated as a source in this case. The color-coded boxes in Figure C.1 refer to Figure C.2 

to Figure C.8, which show that section of the model in more detail.

Figure C.1.—Nominal Recovery Model.
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Figure C.2.—Source to Line Fault.
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Figure C.3.—Line Fault and Energy Storage.
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Figure C.4.—L-2 Branch.
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Figure C.5.—Inverter and Motor Connection.
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Figure C.6.—Right Engine Emulation.
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Figure C.7.—L-1 Load Emulation.
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Figure C.8.—L-1 Source and Load Emulation.
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Appendix D.—Strathclyde Report
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