NATIONAL BUREAU OF STANDARDS REPORT

9886

CIRCUIT DESCRIPTION

OF A PROTOTYPE

CABLE FAULT LOCATOR

<NBS>

U.S. DEPARTMENT OF COMMERCE

NATIONAL BUREAU OF STANDARDS
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NATIONAL BUREAU OF STANDARDS REPORT

NBS PROJECT

2120411

NATIONAL BUREAU OF STA

for use within the Government. B and review. For this reason, the whole or in part, is not authoriz Bureau of Standards, Washington

the Report has been specifically p

July 22, 1968

CIRCUIT DESCRIPTION

OF A PROTOTYPE

CABLE FAULT LOCATOR

by

David Brenner

Electronic Instrumentation Section

Measurement Engineering Division

for Photometry Section

Metrology Division

IMPORTANT NOTICE Approved for public release by the

Director of the National Institute of

Standards and Technology (NIST) on October 9, 2015

NBS REPORT

9886

accounting documents intended

ubjected to additional evaluation

isting of this Report, either in Office of the Director, National

the Government agency for which

)ies for its own use.

U.S. DEPARTMENT OF COMMERCE

NATIONAL BUREAU OF STANDARDS

i

ABSTRACT

This report describes the circuitry and electrical performance of a prototype Cable Fault Locator test set. The primary function of the test

set is to locate the position of faults in single conductor, unshielded,

buried cable such as that used in series-string lighting installations.

The test set consists of two major parts: an eleven-watt audio-frequency

signal generator which is used to supply a test current to the faulted

cable, and a hand-carried detector unit which is used to measure the

relative strength of the magnetic field produced by the test current.

If, for example, the fault is known to be a short to ground, it is located

by carrying the detector unit along the cable route and looking for the

place where the magnetic field drops sharply.

The test set described is an updated version of a test set produced

by Texas Instruments Incorporated in 1955. The new test set is fully

transistorized and is well protected from overloads and operator errors.

The signal generator supplies a square-wave signal at either 150, 270, or

570 Hz. The operator can select either a continuous test signal; or, as an aid to identifying the test signal under poor signal conditions, an

interrupted test signal can be chosen. A multi ply-tapped output transformer

is used to match the signal generator to its load. The detector unit is

a three-band tuned amplifier. It has a "Q" of 13, maximum voltage gain of

250 000, is powered by four 1.5 volt penlite cells, and drives both

headphones and an indicating meter.
PREFACE

This report is written primarily for the reader who is interested in

the circuitry used in the Cable Fault Locator test set, but attention is

also given to the requirements of those who operate and maintain the

equipment. Section I contains background information about the uses and

features of the equipment, and should be of interest to anyone having

contact with the test set. The information contained in Section I and in

the Controls and Characteristics portion of Sections II and III constitutes

an informal set of operating instructions. Sections II and III contain

the detailed circuit descriptions of the Test Signal Oscillator and of

the Detector Unit, respectively. These two sections are written primarily

for the circuit designer, but are believed to be readable enough to be

helpful to repair personnel. The waveform and voltage measurements,

performance data, and parts lists should be especially useful to the repairman.
TABLE OF CONTENTS

Abstract i

Preface . ii

1. Introduction 1-1

2. Test Signal Oscillator

Controls and Characteristics ..... 2-2

Design Considerations . 2-3

Circuit Description 2-4

power supply 2-4

oscillator 2-7

frequency-shift multivibrator 2-9

frequency divider 2-12

driver circuit 2-12

power amplifier 2-13

Circuit Diagrams 2-15

Description of Circuit Components 2-19

Waveforms 2-21

3. Detector Unit

Controls and Characteristics 3-2

Design Considerations 3-3

Circuit Description 3-4

probe 3-4

tuned input amplifier 3-4

intermediate amplifier 3-11

power amplifier 3-14

operational rectifier and meter 3-15

power supply 3-17

Test Data 3-19

Circuit Diagram 3-23

Description of Circuit Components . 3-25

4. Miscellaneous

Improvements Suggested for the Cable Fault Locator 4-2

References 4-3

Section 1

INTRODUCTION
1-2

FIGURE

1-1.

Photograph

of

the

front

panel

of

the

Test

Signal

Oscillator.
1-3

FIGURE

1-2.

Photograph

of

the

interior

of

the

Test

Signal

Oscillator.
SIGNAL

STRENGTH

1-4 \
1-5

FIGURE

1-4.

Photograph

of

the

printed

circuit

board

and

interior

of

the

Detector

Unit.
1-6

INTRODUCTION

The Cable Fault Locator described in this report is a two unit test set

whose principal use is to locate the position of faults in single-conductor

buried electric cable. The two units which comprise the Cable Fault Locator

(CFL) are the Test Signal Oscillator (TSO) and the Detector Unit (DU).

Photographs of these units are shown in Figures 1-1, 1-2, 1-3, and 1-4.

Figure 1-5 illustrates the technique used to locate the position of a cable

fault, in this case a short to ground. The cable under test is disconnected

from its normal power source and connected to the TSO instead. The TSO

supplies a test current of known frequency to the cable. The DU senses the

magnetic field produced by the test current and provides visual (meter) and

audible (headphones) outputs simultaneously. The DU is carried along the

cable route and indicates a drop in signal strength as it passes beyond the

fault. Since the DU senses the magnetic field produced by the test current,

any situation which reduces the magnetic field intensity available to the DU

reduces the effectiveness of the test procedure. For example, if the cable

is in a metal conduit, has metal armor, or has its return circuit nearby,

then the magnetic field intensity is reduced. Under favorable conditions the

CFL is expected to locate a cable fault to ground to within two feet of the

fault, while an open-circuit type of fault would be located to within approximately twenty feet of the fault. A comprehensive guide to the use of the CFL

is: "Guide to Use of the AN/TSM-11 Cable Test-Detecting Set".'

1' The techniques

of fault location which were developed for the AN/TSM-11 Test-Detecting Set

are valid also for the CFL.

The CFL is a prototype for an improved version of the AN/TSM-11 Cable

Test-Detecting Set which was built in 1955 by Texas Instruments Incorporated.

Most of the improvements incorporated into the new design were made either at

the suggestion of those who had used the earlier units in the field, or because

improved electronic components and circuits have become available. Figure 1-6

lists some of the principal improvements provided by the CFL.
1-7

FIGURE

1-5

,

How

Cable

Fault

Locator

test

set

is

used

to

locate

position

of

a

short

to

ground
1-8

1

Item AN/TSM-11 CFL

Test Signal Oscillator power output, max 6 watts approximately 11 watt

Test Signal Oscillator voltage output, max 300 V rms 600 V rms

Test Signal Oscillator output metered no yes

Detector Unit voltage gain, max 200,000 250,000

"Q" of Detector Unit 6 12.7

Detector Unit response, meter reading

versus input signal amplitude

non-linear linear

Battery type used in Detector Unit special 1.5V penlite cells

Detector Unit battery voltage metered no yes

Incorrect battery polarity applied to

Test Signal Oscillator or to Detector

Unit

damage

possible

no damage

Test frequencies 250 Hz 150, 270, 570 Hz

Figure 1-6. Electrical characteristics of the new Cable Fault Locator (CFL)

compared to the characteristics of the old AN/TSM-11 test set.
1-9

The CFL can be operated at any one of three test frequencies, instead of

just one frequency as was the case for the old test set. All three frequencies

j are selected to be as far away as possible from 60 Hz power line harmonics.

The three frequencies used are 150, 270, and 570 Hz. The test frequency desired

is selected by a front panel switch on the TSO. The DU is tuned to the same

frequency by a similar switch on its front panel. The 150 Hz frequency was

chosen as the lowest frequency likely to be useful. A low testing frequency

has the advantage that there is relatively little coupling of the test signal

into cables which lie adjacent to, or cross, the cable under test. The

disadvantages of the 150 Hz test signal are that the adjacent power frequency

harmonics are relatively strong, the induced voltage in the magnetic pickup

probe of the DU is weak because the rate of change of the magnetic field is

low, and the human ear is not very sensitive to low-frequency signals (see ; -. Figure 1-7). The 570 Hz frequency is the most sensitive in all respects. The

power-frequency harmonics are weaker near this frequency, the human ear is -

more sensitive, and more voltage is induced in the pickup probe for a given

^current in the cable under test. The 570 Hz frequency probably gives the

best results in locating open-circuit faults because the higher frequency

produces larger capacitive currents. The 270 Hz test frequency has characteristics which lie between those of the 150 Hz and 570 Hz frequencies, and is

close to the frequency used in the old test set. The use of three frequencies

was suggested by those who had used Vha j oid test set in the field, and the

usefulness of multiple frequencies can be decided only by field testing of the

new test set.

The CFL is able to employ either unmodulated or modulated test signals.

The DURATION OF FREQUENCY SHIFT IN MILLISECONDS switch on the front panel of

the TSO selects one of three modulation modes: l) continuous uninterrupted

test signal; 2) test signal interrupted for 20 ms approximately 130 times per

minute; 3) test signal interrupted for 200 ms approximately 95 times per minute.

The interrupted signals help the operator to distinguish the test signal from

background noise and interference. The 20 ms interruption is expected to be

preferable for most purposes because it enables the test signal to be identified
Relative

intensity

in

Decibels

1-10

FIGURE * Threshold of hearing for average listener . The average listener

can hear sounds whose intensity lies above the curve. The graph

is based on Figure 14.4 of Reference 2.

I
1-11

by both ear and eye. The ear can hear the interruption and the eye can detect

the resulting small dip in the meter reading. The dip in the meter reading

is small enough so that the meter is still easily read. The ear is able to

detect signals which are far too weak to be detected by the meter. When the

signal is very weak or when the signal-to-noise ratio is low, using the 200 ms interruption should help the ear to detect and identify the test signal. The

uninterrupted test signal is useful when measuring the output frequency of

the TSO.

The CFL instruments are designed to be operated and stored at temperatures

between -20°C and +60°C.
‘ I

. . . 4 i
2-1

Section 2

TEST SIGNAL OSCILLATOR

The Test Signal Oscillator (TSO) is a power oscillator which supplies

a square-wave output signal. The frequency, voltage, and modulation of the

output signal are selected by front panel switches. Approximately 11 watts

is delivered to matched loads. A multiple-tap output transformer matches the

TSO to its load. Selection of the optimum tap is quickly accomplished by

observing the indication of a front panel meter.
2-2

CONTROLS and CHARACTERISTICS

The TSO may be powered from either an ac source of 110-130 V rms at

55-65 Hz, or from 10-16 Vdc. The maximum current drawn from an ac power

source is 0.4 A rms, or 1.9 A dc if a dc power source is used. Two input

power connectors, one for ac and one for dc, are mounted on the front panel.

The TSO is protected from incorrect polarity of the dc power source by a series diode. It is permissible to have both the ac and the dc power sources

connecxed simultaneously. The pilot lamp is connected to an unregulated

power supply in the TSO, so the lamp brightness can be used to estimate the

voltage of the power source. The ground wire of the ac power input is the

only connection made to the case of the TSO.

CIRCUIT BREAKERS The ON-OFF switch is also the reset switch for three

mechanically ganged but electrically independent circuit

breakers. If any one of the breakers should trip, the

other two will also open. The circuit breakers are reset

by turning the switch to OFF and then back to ON.

FREQUENCY

IN HZ

This switch selects the frequency of the test signal at the

SIGNAL OUTPUT connector. The test frequencies are 150, 270,

and 570 Hz. Frequency tolerance is ±2$.

DURATION OF

FREQUENCY SHIFT

IN MILLISECONDS

This switch selects one of three test modes: continuous

single -frequency signal; test signal interrupted for 20 ms

approximately 130 times per minute; or test signal interrupt^

for 200 ms approximately 95 times per minute. The interrupt!

is accomplished by momentarily increasing the test signal

frequency to roughly 1.7 times its nominal value.

OUTPUT VOLTAGE

IN VOLTS RMS

This switch determines the nominal open-circuit voltage at

the SIGNAL OUTPUT terminals by selecting the appropriate tap

on output transformer T2. The nominal voltages are 0, 6, 13,

28, 60, 130, 280, and 600 volts rms, when the power source

is either 130 volts ac or 15 volts dc. Maximum power into

the load is achieved by selecting the switch position which

provides the largest on-scale reading of the POWER AMPLIFIER

COLLECTOR CURRENT meter.

SIGNAL

OUTPUT

The test signal supplied by these terminals is a square wave

whose frequency, modulation, and amplitude are determined by

the three controls described immediately above. Any passive

load may be connected to these terminals. The TSO is protect

from short-circuiting of these terminals. However, the TSO

may be damaged if more than six volts ac or dc is developed

across these terminals by an external source. The black

terminal is the circuit ground and is isolated from the

instrument case.

POWER AMPLIFIER

COLLECTOR CURRENT

This ammeter measures the total dc current drawn by the Powei

Amplifier stage (Q17-Q20). The Power Amplifier is operating

within its design rating as long as this meter reads on-scal"

An off-scale reading does not endanger the Power Amplifier bu

may cause the circuit breakers to trip.
2-3

DESIGN CONSIDERATIONS

To the greatest extent possible the circuits selected for use in the TSO

use standard components, require only one circuit adjustment, and yield easily

to analysis. The transformers and filter chokes are the only components which

were custom built for the TSO. All other electrical components are standard

catalog items. The benefits derived by employing circuits amenable to analysis

are twofold. First, component values can be selected by calculation rather

than trial and error; and second, comparison of calculated performance with

measurements made on the finished circuit provides an excellent method for

detecting both design and construction errors.

Because the TSO will normally be used at military airfields, an attempt

was made to design the circuits with components which are readily available

from military supply sources. To this end, all semiconductor components are

chosen from the ’’Military Standard Preferred and Guidance Lists of Semiconductor

Devices”, MIL-STD-701E , 8 December 1963. Restricting the semiconductor

complement to this list did not cause any serious circuit design problems,

although it did increase the cost of the transistors and require using four

power transistors in the Power Amplifier stage where two non-military transistors

would perform as well. It was not possible to obtain delivery in a reasonable

time for the military quality semiconductors, so ordinary commercial units with

the same device number are employed in their place.

The packaging selected for the TSO is simple but sturdy. The layout of

both the chassis and the printed circuit board is roomy and all components

are reasonably accessible. The front panel is water tight. The electrical

connectors and the panel meter are sealed to the panel with silicone adhesive

and probably cannot be removed without destroying this seal. The pilot light,

switches, and machine screws are sealed by gaskets and can be easily removed

and re-used if required. On the other hand, none of the joints or seams of the

protective aluminum case have been made water tight.
2-4

CIRCUIT DESCRIPTION i

A block diagram of the Test Signal Oscillator (TSO) is shown in Figure 2-1,

The Power Supply section provides unregulated dc voltage to the Power Amplifier

and regulated 6.8 Y dc to the rest of the instrument. The Power Supply can be

energized by either a 110-130 V ac line or by a 10-16 volt battery or dc

generator. The Oscillator section operates at twice the frequency appearing

at the instrument’s SIGNAL OUTPUT terminals. The Oscillator feeds the

Frequency Divider which divides the oscillator frequency by two. The Frequency

Divider feeds the Driver stage which in turn supplies the required base

current drive to the Power Amplifier. The Power Amplifier is a push-pull

class B amplifier and uses an output transformer with multiple secondary

taps to provide impedance matching to the load. The Frequency Shift

Multivibrator is a free-running multivibrator which periodically turns on

transistor Q7, causing the frequency of the Oscillator to increase.

Power Supply (Ql. Q2 . Q.3) : " The Power Supply circuitry (shown in Figure 2-3) consists of a center-tapped

full -wave rectifier, a choke input LC filter, and a voltage regulator. Three

magnetic circuit breakers, ganged to each other and to the power (ON-OFF) swixch,

provide overload protection. Voltage for the pilot lamp and the Power Amplifier

is taken directly from the output of the LC filter while all the remaining

circuits are powered by the 6.8 volts supplied by the voltage regulator.

The voltage regulator is composed of transistors Ql, Q2, and Q3. In order

to enable the regulator circuit to function when the voltage drop across pass

transistor Q3 is low, constant-current generator Ql, rather than a resistor,

is used to provide the base drive to Q3. CIO, shown in Figure 2-4, supplies

transient switching current to the Driver stage (Q13, Ql6*). The voltage

regulator has been observed to oscillate' under certain load ' conditions when

CIO was between 0.02 and 2.0 pF.

f
2-5

Figure

2-1.

Block

diagram

of

Test

Signal

Oscillator.
2-6

The Power Supply may be energized by either 110-130 volts at 55-65 Hz,

or by 10-16 volts de. Although the TSO requires that the ac and the dc power

sources be connected to the proper input terminals, it is permissible to

have both power sources connected simultaneously. When both ac and dc power

are applied, the TSO draws power from the source providing the higher voltage

at the collector of Q3. When the ac supply happens to develop the higher

voltage, CR4 prevents current from entering the dc source. If polarity is

accidently reversed when connecting to a dc power source, the blocking action

of CR4 will prevent damage to the TSO.

When a dc power source is used, the initial charging current drawn by

Cl might trip the 2.0 ampere breaker if CR3 were not present to prevent the

current surge. Diode CR3 is shunted by R4 in order to maintain voltage on

Cl when a dc power source is used. Aluminum electrolytic capacitors last

longer when stored with a bias voltage than without it.

The thermistor in series with the primary of the power transformer is

required to prevent the 0.4 ampere breaker from being tripped by the turnon current transient. Without the thermistor the transformer would saturate

and draw a large current transient when the power switch is turned on, unless

the switch happened to be turned on near a peak of the ac line voltage waveform

It is believed that the TSO can safely withstand the application of power

voltages up to twice the maximum rated values, although no tests were conducted

to verify this belief. Whenever the input supply voltage is high enough to

make the voltage at the collector of Q3 rise above IS volts, the zener diodes

(CR14, CR15) in the Power Amplifier circuit will conduct. If the excessive

voltage is supplied by a dc source, then the current drawn by the zener diodes

will trip the 1.6 ampere breaker. When the contacts of the 1.6 ampere breaker

open, CR16 provides a conducting path for the inductive current in L2 and T2,

thereby protecting the contacts of the 1.6 ampere breaker. If the excessive

voltage is supplied by an ac source, then saturation of the iron core in Tl,

and/or the load imposed by zener diodes CR14 and CR15 will trip the 0.4 ampere

breaker

.

<
2-7

Oscillator (Q4 - Q10) : A block diagram of the oscillator is shown in Figure 2-2. The oscillation

cycle begins with C2 charging toward the 6.8 volt supply voltage. When the

voltage across C2 reaches one-half the supply voltage, the one-shot is triggered

via Q5 and QS. The one-shot turns on reset transistor Q4 for 40 ps, discharging

C2 to ground. When Q4 turns off, C2 again charges toward the supply voltage

and the cycle repeats.

The oscillator frequency, f^, is twice the frequency appearing at the output

of the TSO and can be calculated from the equation

fn

= 12^ Hz (1)

0.693RC + 40

where R = the value in ohms of Rll, R12, or R13

C = the value of C2 in pF (=0.22 pF).

The quantity 0.693RC is the time interval required to charge C2 to

one-half the supply voltage, and the number 40 is the time interval (in ps)

used to discharge C2. The derivation of Equation 1 is based on the conditions

that C2 charges to exactly one-half the supply voltage and is then discharged

completely to ground. In the circuit actually used, however, tolerance errors

in RIB, R19, and C2; voltage offset between the bases of Q5a and Q5b; and

failure of C2 to discharge fully to ground may cause the actual frequency

generated to differ from the calculated value by several percent. The frequency

is adjusted to the desired (calculated) value by placing a trimming resistoT,

R17, in parallel with either RIB or R19, as required (see Figure 2-4).

In order to ensure good frequency stability the RC timing circuit

(C2; Rll or R12 or R13) is designed so that at least 99% of the current in

the charging resistor is used to charge C2. The emitter junction of Q5a is

reversed biased during most of the charging interval, so the transistor

specified should have no more than 3 pA leakage at room temperature with 3.5

volts base-to-emitter bias. As much as 10 pA base drive may be required by

Q5a in order to initiate the discharge cycle. But currents of this magnitude

are drawn only during the last 50 mV of capacitor charging and thus do not

significantly affect the overall charging time.
-+6.8V

2-8

o

6

w.o

o

o

CO o

CM c

i E

o»o

CM

09

v. .2* IS

Li. O

CD

= Jr^i ^r!Q

oc ®(t 0 o:

>

o

£ l

o

+

40us
2-9

A constant-current source, Q6, is used to supply emitter current to Q5

in order to make the oscillator circuit start at low Power Supply voltages.

The oscillator frequency can be electronically shifted by turning on Q7, as

is described in the section dealing with the Frequency Shift Multivibrator.

The one-shot circuit is used to drive the Frequency Divider circuit as

well as transistor Q4. When the one-shot is in its quiescent state, Q9 is

off and Q10 is conducting, thus keeping 04 biased off. The one-shot switches

state when 09 is driven into conduction by Q8. The rapid fall of the Q9

collector voltage is used to trigger the Frequency Divider circuit, while the

increasing Q10 collector voltage turns on Q4. The one-shot remains in its

switohed condition for 40 ps. The collector voltage of 09 is limited to

around 4.5 volts in order to prevent the emitter-base junction of Q10 from

receiving excessive reverse voltage via C3.

Care was taken to ensure that the one-shot would trigger reliably, since

even one failure to respond to a trigger signal from 08 would prevent 04 from

discharging C2 and the oscillator would hang up. If poorly designed, the oneshot could fail to respond to a slowly rising trigger from 08 because the

presence of C3 in its regenerative feedback loop would block the feedback signal.

In the circuit used here regeneration is assured for any trigger signal by

making the high-frequency loop gain of the one-shot large enough to ensure

spontaneous regeneration when 09 is brought into conduction. Shunting feedback

resistor R25 with a capacitor, and not driving Q10 too far into saturation in

its quiescent state assure the necessary high-frequency loop gain.

Frequency- Shi ft Multivibrator (Oil. 012) : Either a continuous or an interrupted output signal is available from

the TS0. The interrupted signal is obtained by momentarily shifting the

signal to a higher frequency. It is the function of the Frequen -onift

Multivibrator to establish the duration and repetition rate of the interruption.

The frequency shift is accomplished by permitting the Frequency-Shift Multivibrator

to free-run and periodically drive Q7 into conduction (see Figure 2-4). When

Q7 conducts, the base voltage of Q5b drops and causes the Oscillator frequency

}

to increase. A single-frequency, uninterrupted output signal is obtained by

disabling the multivibrator, causing Q7 to remain off.
2-10

The circuitry of the Frequency-Shift Multivibrator is very similar to

that of an ordinary free -running multivibrator. When switch S2 is in the 0 ms

position the regenerative feedback path of the multivibrator is broken and both

transistors conduct indefinitely. When conducting, the collector of Qll is

only 0.5 volts above ground and the series chain of CR10, R28, R29 assures that

Q7 is biased off. Tantalum capacitor C5 is reverse biased when the multivibrator

is in this inactive state. CRll is required to limit the magnitude of the

reverse bias voltage to a few tenths of a volt so that C5 will not be damaged.

If CRll were omitted, the bases of both transistors would be approximately 0.7

volts above ground and each collector would be approximately 0.1 volts above

ground, causing a net reverse voltage of 0.6 volts to appear across C5. The

effect of adding CRll to the circuit is to prevent the collector of Qll from

dropping more than about 0.2 volts below the Qll base, which means the reverse

voltage on C5 is also limited to around 0.2 volts. If the Qll collector

voltage tries to drop below the Qll base voltage, CRll will conduct and drain off

the excess base drive current. The solid tantalum capacitor used for C5 may

be able to stand a 0.7 volt reverse bias without ill effect, but an old

^ tantalum capacitor of unknown construction used in the breadboard circuit

developed excessive leakage current with 0.7 volts reverse bias.

When switch S2 is in the 20 ms or the 200 ms position the regenerative

feedback path is no longer broken and the multivibrator spontaneously begins

to free-run. Reliable starting is assured because the loop gain of the

multivibrator is greater than unity when both Qll and Q12 are on. Diode CRll

keeps Qll out of saturation, ensuring that this stage functions as an amplifier

with reasonable gain. Biasing Q12 so that the ratio I^/lg is reasonably high

ensures that Q12 is not so badly saturated that the gain of the Qll stage is

lost by attenuation in Q12.

In order to describe the operation of the free-running multivibrator, let

us assume that Qll is initially on, Q12 is off. Qll is held on by the base

current supplied by R32, and Q12 is held off by a reverse base voltage provided

by C5. While the transistors are in this stage C5 charges slowly toward the

supply voltage through R31. When C5 has charged sufficiently to bring the base

voltage of Q12 to around 0.5 volts above ground, QI2 begins to conduct and
2-11

initiate a regenerative cycle which quickly drives the transistor pair into

their other stable state. The regenerative cycle proceeds as follows: As

Q12 is brought into conduction its collector voltage drops. The drop in

collector voltage is coupled via C6 (or C7) to the base of Qll, causing Qll

to begin to turn off. As Qll turns off, the current in collector resistor

R30 is diverted into C5 and the base of Q12. This current turns Q12 on more

strongly, reinforcing the initial turn-on current and causing further

regeneration. After the transistors are firmly planted in their new state,

current continues to flow in R30 until C5 is fully charged, and C6 (or C7)

gradually charges towards the supply voltage to initiate another regenerative

cycle.

Ideally, the off transistor will remain off until its timing capacitor

has charged sufficiently to bring it back into conduction, at which point

regenerative feedback quickly flips the circuit into its other state. The

oscillogram of the Q12 base voltage displays the ideal situation (see Figure

2-10). Note that regeneration occurs when the base voltage has risen to about

+0.5 volts, which is the voltage required to drive the transistor into conduction.

On the other hand the oscillogram of the Qll base voltage (Figure 2-9) shows

that the transition takes place when the base voltage has risen to only

+0.25 volts, too small a voltage to turn on Qll.

The premature transition is caused by negative -going transients on the

6.8 volt supply line. These transients are generated by the switching of the

Driver circuit (Q13 and Ql6) and are conducted to the base of Q12 via R30

and C5. When the negative-going transient reaches the base of Q12 it momentarily

turns off Q12 and thereby initiates the transition. Q12 is sensitive to

transients because the path (C5, R30, R3l) from its base to the power line has

a much lower impedance for ac signals than for dc signals. Accurate timing

is not required of the Frequency- Shi ft Multivibrator so this premature transition

causes no real problem.
2-12

Frequency Divider (Q,1A. Q.15) : The Frequency Divider circuit is a conventional binary divider whose

output frequency is one-half its input frequency. The frequency of the divider j output is the same as that appearing at the SIGNAL OUTPUT terminals of the TSO.

The circuit triggers reliably when the leading (negative-going) edge of the

input signal from Q9 has a slope of at least 0.8 V/ps and a frequency of less

than 6000 Hz.

The Frequency Divider circuit switches state each time it receives a negative -going input pulse. In order to describe the switching process, let

us assume that Q14 is initially conducting and that Q15 is off (see Figure 2-4).

The collector voltage of Q15 is high, causing CR13 to be strongly reverse

biased. The reverse-biased diode isolates the base of Q15 from input signals

of either polarity. On the other hand the collector of Q14 is near ground

potential, causing CR12 to be slightly forward biased. CR12 blocks positivegoing input signals, as was the case with CR13, but allows a negative-going

input voltage to reach the base of Q14. When a negative -going voltage arrives

at the junction of C8 and C9, CR12 is driven into conduction and Q14 turns off. I As Q14 turns off, its collector voltage rises and turns on Q15 via R41. As

Q15 turns on, its collector voltage drops and removes the base drive to Q14,

so Q14 remains off after the trigger signal is gone. The transistors remain

in their new state until the next trigger signal arrives. The next trigger

signal must not arrive, however, until C8 and C9 have had time to at least

partially charge toward their new steady-state voltages.

Driver circuit (Q13. Ql6) : The Driver circuit (see Figure 2-4) provides a push-pull drive signal to

the Power Amplifier. The driver transistors are used as switches and are

driven by the Divider circuit. Normally, Q13 will be off when Ql6 is on, and

vice versa. But during the switching transition the transistors momentarily

conduct simultaneously. Stored charge in the base region of the conducting

transistor prevents it from turning off as promptly as the ’’off” transistor

turns on, even though both transistors receive their input signal at the same

instant. The additional power supply current drawn during the interval of

simultaneous conduction is supplied by CIO. I
2-13

Power Amplifier (QJ.7, Q,l8, Q19, 020) : The Power Amplifier is a push-pull class B square-wave amplifier which

drives its load via multiple-tap output transformer T2. The circuit diagram

for the Power Amplifier is shown in Figure 2-4. Optimum matching of the

Power Amplifier to its load is obtained by selecting the tap (switch S3)

which produces the largest on-scale reading of the 0-1.5 ampere meter. The

amplifier is fully protected from overloads. If the dc current drawn by the

amplifier significantly exceeds the design maximum value of 1.5 amperes for

any reason, such as short-circuited SIGNAL OUTPUT terminals, the 1.6 ampere

circuit breaker will trip. If the voltage of the source which powers the

TSO is excessive, zener diodes CR14 and CR15 will conduct during the positive

peak of the square wave and again cause the 1.6 ampere breaker to trip. The

primary purpose of the zener diodes, however, is to clip any voltage spikes

which may appear across the power transistors during normal operation. The

four power transistors and the two 50 watt zener diodes are mounted on a heat sink which is large enough to ensure that the circuit breaker will trip

before any of these components overheat.

The 2 ohm resistor, R51, in series with the secondary of the output

transformer enables the Power Amplifier to drive a short-circuit load from

the 6 volt transformer tap. When a higher voltage tap is used a short circuit

will overload the amplifier. As mentioned in the preceding paragraph, shortcircuit current drawn by the external load is expected to cause the dc current

drawn by the Power Amplifier to become large enough to trip the 1.6 ampere

circuit breaker. The 1.6 ampere circuit breaker is guaranteed to trip at

2.0 amperes, so the amplifier must be designed to draw at least 2.0 amperes

under shortrcircuit conditions. Because rather large current and voltage drive

to the base is required to ensure obtaining two amperes collector current in a single 2N1490 transistor, the power transistors are operated in pairs. The

increased current gain and reduced base voltage drive obtained by dividing the'

load current markedly simplifies the Driver circuit. The power handled by the

Power Amplifier is low enough so that only one transistor per side is required

to handle the power requirements, hence it is unnecessary to force the load

current to divide equally between the two paralleled transistors.
2-14

It is worth noting that the use of dc coupling in the Frequency Divider - Driver - Power Amplifier chain means that at any given time one side of the

push-pull Power Amplifier is conducting. If for any reason the conducting

state fails to alternate between the push-pull halves, the resulting hang-up

will quickly cause the primary of T2 to saturate and short-circuit the Power

Supply. Two details in the circuitry of the TSO prevent this characteristic

from causing trouble. The first circuit detail is the 1.6 ampere circuit

breaker in series with the primary of T2, which protects the TSO from damage

if the Power Amplifier should hang up for any reason. Hang-up would result

if the Oscillator circuitry should fail to oscillate, or from any other

malfunction which disables the push-pull base drive to the Power Amplifier.

The second circuit detail is concerned with obtaining reliable starting of

the TSO when its power switch is first turned on. The potential trouble

here is the possibility of the Power Supply voltage rising so slowly that

the Power Amplifier will drive T2 into saturation before the Oscillator

and the Frequency Divider circuits can start up and provide the required

push-pull drive to the Power Amplifier. For this reason the TSO is designed J so that higher Power Supply voltage is required to bring the Driver and

Power Amplifier circuits into conduction than is required to bring the

Oscillator and Frequency Divider circuits into operation, thus assuring

satisfactory turn-on of the TSO. However, under normal circumstances the

6.8 volt supply line rises to full voltage in less than 0.02 seconds, which

is fast enough to obtain reliable instrument turn-on regardless of the order

in which the various circuits become operational. In other words, when operated

from low voltages T2 cannot saturate in only 0.02 seconds. A switch with

make-before-break contacts is used for SI in order to prevent the Oscillator Circ

from stopping during switching.

As was the case in the Driver circuit, stored charge in the base of the

"on" transistors causes momentary simultaneous conduction of all four transistors

in the Power Amplifier during the switching transition. This simultaneous

conduction would momentarily short-circuit the Power Supply if inductor L2 were

not present. High current pulses are undesirable because they may trip the

1.6 ampere circuit breaker and also make life more strenuous for the transistors.
Notes .:

l) The components which are mounted on the printed circuit board are CR5, CR6,

CR7, Ql, Q2, R3, R5, R6, R7, R8, R9.

FIGURE 2-3

POWER SUPPLY

OF

TEST SIGNAL OSCILLATOR

2) The dc voltages are measured with respect to the circuit common. The SIGNAL

OUTPUT terminals are open-circuited and a 120V, 60 Hz power source is used.

-n_f

SEE FI6 2-16,2-19

-FREQUENCY - SHIFT MULTIVIBRATOR POiER AMPLIFIER

Notes :

l) Switches shown in their extreme ccw position as viewed from the front panel.

2) Components which are not mounted on the printed-circuit board include CR14, CR15,

Q17, Ql8, Q19, Q20, and R51.

TEST SIGNAL OSCILLATOR

/SEE FIGURE 2-3 FOR \

V POIER SUPPLY CIRCUIT'

FIGURE 2-4

2-19

DESCRIPTION OF CIRCUIT COMPONENTS

Test Signal Oscillator

Ammeter

The ammeter is a Honeywell model HS2X, 0-1.5 ampere dc meter. This is

a 2-l/2 M unit which is ruggedized and hermetically sealed.

Capacitors

Cl is an aluminum electrolytic capacitor, Mallory type CG63U25K1. All

other polarized capacitors are tantalum electrolytic with voltage

ratings of 15 or 20 V dc from the Sprague 150D series (Military CS13

series )

C2 is a polycarbonate capacitor, Component Research Company, Inc. part

number 05PL224D. All other unpolarized capacitors are Sprague series

192P ’'Pacer” units having ±10$ tolerance and voltage rating of either

80 or 200 Vdc.

Circuit Breaker Assembly

The circuit breaker assembly is manufactured by Airpax Electronics, Inc.,

Cambridge Division, Cambridge, Maryland. The assembly consists of three

electrically independent breakers which are mechanically ganged to each

other and to the toggle switch. The assembly is a member of the Airpax

AP112 family of electromagnetic circuit protectors, with the three electrically independent poles specified as follows:

Pole 1 1.6 amperes dc Delay 1

Pole 2 2.0 amperes dc Delay 1

Pole 3 0.4 amperes 60 Hz Delay 1

Resistors

Unless otherwise noted, resistors are ±5$ tolerance carbon composition with

0.5 watt power rating.

Resistors which are specified by the circuit diagram to have ±1$

tolerance also have temperature coefficients no greater than 50 ppm/°C.

Commercial units which are suitable include Corning Electronic Components

types NC4 and NC5, and IRC type MEA-T-2

R3 is a disk thermistor, 50 ohms at 25°C, 0.4” diameter, Fenwa'i LB15J1.

R10 is a power resistor from Ohmite 995-5B series.

R51 is a power resistor from the Dale HG-25 series.
2-20

Switches

51 is a shorting (make-before-break) rotary switch.

52 may be either a shorting or a non-shorting rotary switch.

53 is a heavy-duty rotary switch with non-shorting contacts, Centralab

catalog number JV-9033.

Transformers and Chokes

All transformers and chokes were custom made to the following

specificat ions

LI Filter choke, 40 mH at 1.8 A dc, dc resistance less than

1 ohm. Approximate body size is 2.7 x 1.9 x 2.3" high.

L2 Iron core choke, 0.5 mH at 1.5 A dc, dc resistance less

than 0.2 ohms. Approximate body size is 1.4 x 1.1 x 1.2" high.

T1 Power transformer, primary to operate at 130 V rms and 55 Hz,

secondary to supply 18 V rms on each side of center tap at

1.8 A rms. Approximate body size is 3.1 x 2.3 x 2.6" high.

T2 Push-pull output transformer for square waves between 150 Hz

and 570 Hz. Secondary to have taps to provide open circuit

voltages of 6, 13 , 28, 60, 130, 280, 600 V rms when the power

supply voltage at the center tap of the primary is 14 V dc.

The voltage at each tap is permitted to drop 20$ when loaded

so that the primary draws 1.5 A dc. Good square-wave shape

is not required. Approximate body size is 2.8 x 2.5 x 3.4" high.

The above items should be able to operate continuously at 75 °C ambient.

Miscellaneous

The sockets used for power transistors Q3, Q17, Ql8, Q19, Q20 are Augat

Inc. part number 8038-1G1.

The sockets used for zener diodes CR14 and CR15 are Robinson Nugent part

number NS-502-8.
2-21

WAVEFORMS

The oscillograms on the following four pages are the voltage waveforms

at selected points in the TSO. The oscillograms supplement the preceding

circuit descriptions and also should be helpful in trouble-shooting.

Test conditions which apply to all oscillograms are:

FREQUENCY IN HERTZ switch set at 570 Hz.

Power source is 120 volts at 60 Hz.

Vertical deflection labeling is in volts.

Notations used in describing the test conditions of the individual

oscillograms are:

52

refers to the setting of the DURATION OF

FREQUENCY SHIFT IN MILLISECONDS switch.

53

refers to the setting of the OUTPUT VOLTAGE IN

VOLTS RMS switch.

LOAD means the load connected to the SIGNAL OUTPUT

terminals. A 99 ohm load causes full-scale

deflection of the POWER AMPLIFIER COLLECTOR

CURRENT meter.

SYNC — Qll . . . means that the horizontal sweep of the

oscilloscope is triggered by the negative-going

transition at the collector of Qll.

SYNC — Q13 . . . means that the horizontal sweep is triggered by

the positive-going transition at the collector

of Q13.

SWEEP. ...... means the sweep speed of the oscilloscope

horizontal deflection.
2-22

FIGURE 2-5

4

WAVEFORM at base of Q5a

S2 0 ms

2 S3 any

LOAD — any

SYNC — Q13

SWEEP - 200|is/cm

0

FIGURE 2-6

WAVEFORM at collector of Q9

52

0 ms

5

3

any

LOAD — any

SYNC — Q13

SWEEP - 200|is/cm

FIGURE 2-7

4

WAVEFORM at collector of Q10

S2 0 ms

2 S3 any

LOAD — any

SYNC — Q13

SWEEP - 200|is/cm

0
2-23

FIGURE 2-8

4

WAVEFORM at collector of Qll

S2 200 ms

2 S3 any

LOAD — any

SYNC — Qll

SWEEP - O.ls/cm

0

0

-2

-4

FIGURE 2-9

WAVEFORM at base of Qll

52 200 ms

53

any

LOAD — any

SYNC — Qll

SWEEP - O.ls/cm

0

-2

-4

FIGURE 2-10

WAVEFORM at base of Q12

52 200 ms

53

any

LOAD — any

SYNC — Qll

SWEEP - O.ls/cm
2-24

FIGURE 2-12

8

WAVEFORM at collector of Q13

S2 0 ms

4 S3 60 volts

LOAD — not known

SYNC — Q13

SWEEP - 200ns/cm

0

FIGURE 2-13

8

WAVEFORM at collector of Ql6

S2 0 ms

4 S3 60 volts

LOAD — not known

SYNC — Q13

SWEEP - 200|is/cm

0

FIGURE 2-14

40

WAVEFORM at collector of Q20

S2 0 ms

20 S3 60 volts

LOAD — 99 ohms

SYNC — Q13

SWEEP - 200M.s/cm

0

FIGURE 2-13

WAVEFORM at collector of Q17

S2 0 ms

20 S3 60 volts

LOAD — 99 ohms

SYNC — Q13

SWEEP - 200[is/Cm
2-25

FIGURE 2-16

+ 40

WAVEFORM at SIGNAL OUTPUT terminal

S2 0 ms

0 S3 60 volts

LOAD — 99 ohms

SYNC — not known

SWEEP - 200n.s/cm

-40

FIGURE 2-17

WAVEFORM at collector of Q20

52

0 ms

53

60 volts

LOAD — open circuit

SYNC — Q13

SWEEP - 200|is/cm

FIGURE 2-18

WAVEFORM at collector of Q17

52 —— 0 ms

53

60 volts

LOAD — open circuit

SYNC — Q13

SWEEP - 200^.s/cm

+ 100

FIGURE 2-19

WAVEFORM at SIGNAL OUTPUT terminal

S2 0 ms

0 S3 60 volts

LOAD — open circuit

SYNC — not known

SWEEP - 200|is/cm

-100

3-1

Section 3

DETECTOR UNIT

The Detector Unit (DU) is a hand-carried magnetic field monitor. Voltage

induced in its pickup probe by fluctuating magnetic fields drives a tuned

amplifier which in turn drives both an indicating meter and headphones. The

amplifier passband and voltage gain are selected by front panel controls.
3-2

OFF

150 Hz

270 Hz

570 Hz

BATTERY

TEST

GAIN

Meter

HEADPHONES

BATTERIES

CONTROLS and CHARACTERISTICS

When the rotary switch is in this position the battery pack

is disconnected from the circuit and the DU is inoperative.

These switch positions select the frequency to which the DU

is tuned. The Meter indicates relative signal strength.

This switch position connects the Meter to measure the voltage

of the battery pack. The battery voltage is satisfactory

when the meter reading is at least 70$ of full scale (i.e.

battery pack voltage at least 4.5 volts). Except for the*

change in the Meter function the DU circuits operate as

usual as a 570 Hz tuned amplifier.

This control varies the voltage gain from the PROBE input

connector to the HEADPHONES jack from approximately 250 000

down to less than 600 when a 500 ohm load is used. For a given setting of the GAIN control the meter reading is

proportional to the signal voltage induced in the Probe,

provided the GAIN control pointer is within the wide, white

circular arc marked on the panel. For lower gain settings

the DU may saturate and cause the meter indication to be

nonlinear.

^ The Meter indicates either relative signal strength (see

GAIN control, above) or battery voltage, depending on the

position of the rotary switch. The meter reading is not

affected by loading of the HEADPHONES jack.

The signal available at this jack is the amplified Probe

voltage. Four milliwatts is delivered to a 500 ohm load

when the signal- strength meter reads full scale. 500

ohms is the preferred load impedance but any other load,

including a short circuit, can be safetly driven from

this jack. Output impedance is roughly 350 ohms.

Access to the four penlite cells is obtained by unscrewing

the threaded insert which is located at one end of the case

Any type of ”AA" size cell (9/l6” dia x 1-31/32" long) which

supplies betv/een 1.1 and 1.7 volts can be used. Carbon—zinc

cells designed for use in transistor radios (Eveready #1015 or equivalent) are recommended for use at ambient temperatures

between -15 and + 40°C (+5 to +104°F).

#
3-3

DESIGN CONSIDERATIONS

The circuit design of the Detector Unit follows the same guidelines used

for the Test Signal Oscillator. The DU requires no circuit adjustments, most

of its circuits can be easily analyzed, and except for its pickup Probe standard

components are used. Most of the semiconductor components are silicon, but

germanium semiconductors are used where their smaller base-to-emitter voltage

results in better circuit performance. All diode and transistor types used are

chosen from the "Military Standard Preferred and Guidance Lists of Semiconductor

Devices", MIL-STD-701E, 8 December 1965. Ordinary commercial grade semiconductors

were installed in the DU if military grade units could not be obtained.

The mechanical design of the DU is intended to enhance accessibility of

the circuit components rather than to obtain a very small instrument package.

The frame of the DU is machined from a solid block of 6061-T6 aluminum alloy.

Two removable panels provide access to the circuit components. Two edges of

the printed circuit board rest in a slot milled in the frame. A machine screw

near a third edge holds the board in place. Connecting wires are long enough

to permit the instrument to operate with the printed circuit board removed from

the case. The wiring layout on the circuit board is very similar to the

schematic diagram layout and all components are labeled, hence trouble shooting

is made easier. All components and mounting hardware on the front panel are

watertight. The other fittings and joints are not necessarily watertight.
3-4

CIRCUIT DESCRIPTION 1

A block diagram of the Detector Unit is shown in Figure 3-1. Audiofrequency current in the cable under test produces a magnetic field which

induces a small voltage in the Probe. This voltage is amplified by the

Tuned Input Amplifier. The frequency response of the Tuned Input Amplifier

approximates that of a single-tuned RLC circuit with a "Q" of 12.7. Since

the voltage gain is small for undesired frequencies, saturation of the

amplifier by strong but unwanted pickup from nearby power lines is unlikely.

GAIN control R17 is used to attenuate the output signal from the Tuned

Input Amplifier in order to prevent saturation of the Intermediate Amplifier

which follows. The Intermediate Amplifier provides additional voltage gain

and drives the Power Amplifier and the Operational Rectifier. The Power

Amplifier drives the Headphones. The Operational Rectifier provides full-wave

rectification of the ac signal and drives the Meter.

Probe:

The Probe is a coil of wire wound on a high-permeability rod and mounted

at the end of a telescoping tube. At 1000 Hz the Probe impedance is equivalent

to 5.7 mH in series with a 5 ohm resistor. The Probe assembly is the one

originally used with the old AN/TSM-11 Test Set.

Tuned Input Amplifier (Q.l - Q5) : A block diagram of the Tuned Input Amplifier is shown in Figure 3-2.

The low-pass filter prevents radio-frequency signals from entering the amplifier.

Input transformer T1 improves the signal-to-noise ratio by providing a degree of

matching between the amplifier and the low-impedance Probe. The parallel-T

feedback network produces the frequency-selective characteristic of the amplifier

(see Figure 3-3).

The impedance seen looking into the PROBE connector is approximately

3 mH in series with 5 ohms resistance. This approximation is useful from dc

to 1000 Hz. Radio-frequency signals which may be picked up by the Probe or

its leads are prevented from entering the transistor circuitry by the low-pass

filter. Because the amplifier gain is very low at radio frequencies,

4
Power

3-5

FIGURE

3-1.

Block

diagram

of

Detector

Unit.

Items

within

dotted

boxes

plug

into

the

main

chassis.
3-6

£<c

<*8

Figure

3-2.

Block

diagram

of

Tuned

Input

Amplifier.
3-7

FREQUENCY IN Hz

Figure 3-3. Frequency response of Tuned Input Amplifier.

I
3-8

direct amplification of high-frequency signals is not a source of trouble.

Instead , the amplifier’s sensitivity to rf signals is caused by the fact

that the base-emitter junction of a transistor is normally operated as a forward-biased diode and is able to rectify signals having frequencies of

tens of MHz. The dc and low-frequency components of the rectified signal

constitute undersired signals to which the amplifier can respond.

Input transformer T1 gives a voltage step-up of approximately 6.5.

A higher voltage step-up would improve the signal-to-noise ratio but reduce

the input signal voltage which could be handled without overloading the

amplifier. Since the amplifier has no gain control ahead of it, it must

be able to handle large input signals without saturating. The largest input

signal which will not cause saturation is equal to the maximum output voltage

swing (at Q5 emitter) divided by the voltage gain at the frequency of interest.

The amplifier can handle sine-wave input signals of at least 3 mV rms in its

passband and much larger signals outside of its passband.

The heart of the inverting amplifier is Q3, which is the only transistor

providing voltage gain. The remaining three transistors are necessary to

obtain an amplifier which is compatible with its surrounding circuitry.

Constant-current source Q2 (see circuit diagram, Figure 3-11 ) decouples the

base of Q5 from the dc supply and also increases the signal swing available

at the collector of 03. CR2 functions as a voltage regulator which establishes

the base voltage of Q2. Use of this diode is necessary to make the current

supplied by the Q2 collector insensitive to ripple on the dc supply. Power

supply ripple of 1% causes less than .02$ ripple across CR2. Use of current

sink Q4 increases the output-signal swing available from emitter follower

05. The quiescent voltage at the Q5 emitter is equal to the sum of the voltage

across CR1 and the emitter junctions of Q1 and Q3. The output signal at the

05 emitter is capacitively coupled to the GAIN control R17. If the dc voltage

were not blocked, the Intermediate Amplifier would be subject to a momentary

overload whenever the setting of R17 is changed.

The frequency selective characteristic of the Tuned Input Amplifier is

obtained by placing a parallel-T network in the feedback path of the inverting
3-9

amplifier. The parameters chosen here for the parallel-T produce a network

which ideally has zero transmission at one frequency (see Figure 3-4b). At

this frequency there is very little negative feedback and the closed-loop

gain of the inverting amplifier is equal to its open-loop gain. When the

signal is shifted to either side of this frequency, transmission through the

parallel-T network rapidly increases and the resulting negative feedback

reduces the amplifier gain. The parallel-T network should be driven from a low-impedance source and terminated with a high-impedance load, otherwise

3 the selectivity of the Tuned Input Amplifier will suffer. Emitter followers

Q5 and Q1 provide the required impedance levels.

Although the parallel-T network is the predominant frequency determining

element, phase shift in any other element of the feedback loop also influences

the frequency at which the Tuned Input Amplifier response peaks. Phase-shift

effects are negligible for the 150 Hz and the 270 Hz bands but cause the

amplifier peak for the 570 Hz band to occur about 1.8$ below the null frequency

of the parallel-T network. This undesired phase shift is produced by C4, the

high-frequency roll-off capacitor. The cause of the 1.8$ frequency shift was

not discovered until after the DU had been sent to the Visual Landing Aids

Laboratory for field tests and therefore was not corrected. The suggested

remedy is to change the position of C4 so that it is connected between the

collector of Q3 and the circuit ground. This will reduce the amplifier phase

shift and greatly reduce the frequency shift now present.

If the M Q

M of the selective amplifier is defined as the ratio of the

center frequency and the 3 db bandwidth, then the "Q" is approximately equal

to 0.25A, where A is the magnitude of the loop voltage gain with the parallel-T

network ignored. Hence, A = 60 x 0.85 = 51 and the ”Q" is near 12.7. The

moderate open-loop voltage gain of 51 is doubly desirable because it makes

oscillation unlikely and keeps the frequency response broad enough so that

nominal changes in component values will not seriously alter the passband.
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3-11

The parallel-T type of frequency selective network was chosen because

3 4 5

there is adequate design information available, ' f its components

are physically small, band-switching is not too troublesome, and low

temperature coefficient resistors and capacitors are readily available to

make the network fairly temperature insensitive.

Intermediate Amplifier (Q.6 - Q10) : The Intermediate Amplifier is a dc coupled amplifier utilizing negative

feedback to provide ac gain stabilization and to establish the quiescent

operating point. The circuit diagram of the Intermediate Amplifier is shown

in Figure 3-11. The nominal voltage gain from the arm of R17 to the collector

of Q9 is 400 and is down 10$ at approximately 90 and 2500 Hz.

A large signal swing at the amplifier output is obtained by biasing the

collector of Q9 near one -half of the supply voltage and by employing

constant-current source Q10. The quiescent output voltage is determined by

the voltage established at the base of Q7, since the voltage at the base of

Q6 must adjust itself to very nearly the same value. The resistance of R28

and R30 is low enough to ensure that the dc voltage at the base of Q6 is

within 0.2 volts of the Q9 collector voltage. Q8 may conduct strongly when

power is switched on, so R23 is necessary to limit the Q8 collector current.

C9 serves to reduce the high-frequency loop gain of the amplifier and thus

prevent high-frequency oscillation. R25 and C8 provide power supply decoupling.

Signal voltage at the base of Q8 produces feedback to the bases of input

transistors Q6 and Q7. The desired feedback path is from the base of Q8,

through Q9 and the feedback network, to the base of Q6. This path provides

negative feedback (see Figure 3-5). Undesired feedback to the base of Q7

is caused by the Q8 emitter current flowing in the nonzero power-line

impedance (see Figure 3-6). The voltage developed across the power-line

impedance is coupled through R22 to the base of Q7 and is positive feedback

at low frequencies. Low-frequency oscillation will result unless the negative

feedback signal at the base of Q6 is large enough to override the positive

feedback. Early versions of the amplifier would frequently break into low-
3-12

(

( R

gq

+ l/j~C1Q ) (R

1ft + 1/J»C6

) e

i (R

30

+ R

29

+ 1/JMC

10 ) (R

29

+ R

28

+ R

i8

+ l/j^10 + " (R

29

+ l/jftC

l(

.5 = V=1 o> = 27Tf

The electrical values of the above components are

given in the schematic diagram shown in Figure 3-11.

FIGURE 3-5. Transfer function of the principal feedback path . The above

transfer function is valid if the feedback path terminates at

a high-impedance load. The network is loaded, however, by the

input impedance of Q6. This loading causes the magnitude of

eo/e i to drop to around 0.85 of the open-circuit value above

20 Hz, and to around 0.4 of the open-circuit value below 0.5 Hz.
3-13

base of 08

I

_o = e 22

e

i (Ge

+ °25 + G

22 + >C

3

} (G22

+ G

21 + JwC

7

) “ Gh

j = VT ^ = 2 TTf

The electrical values of the components shown above are

given in the schematic diagram shown in Figure 3- 11. The symbol

G means that the resistor values are to be expressed in mhos

(units of conductance) instead of in ohms. G is the intrinsic

conductance of the Q8 emitter junction. Its value in mhos is

approximately equal to the emitter current, expressed in

milliamperes, divided by 30 ; which is .011 mho for the case at

hand.

Figure 3-6. Simplified circuit of parasitic feedback path . The above simplified

circuit shows the principal elements of the parasitic feedback path.

Although the effects of battery resistance and of loading by the

base of Q7 are omitted, the behavior of the complete circuit is

described without serious error.
3-14

frequency oscillation at roughly 1 Hz when the dc power was applied. The

amplifier could be stabilized by momentarily shorting the output to ground

with a large capacitance. In other words, stability criteria which would

prevent oscillation under steady-state conditions did not necessarily prevent

the circuit from breaking into sustained oscillation when initially turned

on. Since analysis of the steady-state stability criteria was of little use,

values for the circuit parameters which affected stability were established

experimentally. In particular, changing either C7 or CIO by a factor of

ten in either direction does not cause oscillation. Increasing the value of

CIO or decreasing the value of C7 moves the circuit in the direction of

oscillation. A large voltage gain at low frequencies is very desirable in

the Q9 stage, for this increases the magnitude of the negative feedback

signal without increasing the positive feedback signal.

Power Amplifier (Qll, 012) : The Power Amplifier is a push-pull circuit designed to drive 500 ohm

headphones. The amplifier can, however, safely drive any other load impedance,

including a short circuit. It delivers 4 mW into a 500 ohm resistive load

when the signal strength meter reads full-scale. The circuit diagram of the

Power Amplifier is shown in Figure 3-11. The voltage gain of the amplifier

from the collector of Q9 to the HEADPHONES jack is approximately 1.7 when

loaded with 500 ohms.

The voltage developed across R32 and CR5 biases Qll and Q12 slightly

ON when no signal is present. Diode CR5 is used to ensure that satisfactory

bias voltage is maintained in spite of changes in battery voltage and

temperature. The voltage developed across R32 increases the bias voltage

slightly so that the emitter current in Qll and Q12 is large enough to

establish a reasonably low value of small-signal internal emitter resistance.

If the internal emitter resistance is too high, the voltage gain from the

base of Qll (and Q12) to its emitter will suffer. Emitter resistors R34

and R35 ensure that the quiescent emitter current will be satisfactory

without requiring that the base-emitter voltage of Qll and Q12 be individually

matched to the voltage drop 'across' D,5. R34 and R35 differ in resistance in

order to compensate for a difference In the dc resistance of the two halves

of the center-tapped winding of T3. R33 is required to damp out a resonance

of Cll with the inductance of T2 which occurs at approximately 150 Hz.
3-15

Operational Rectifier and Meter (Q.13 - Q15) : The Operational Rectifier and Meter form an ac voltmeter. The meter

reading is proportional to the ac voltage applied to R36 . A block diagram

of the circuit is shown in Figure 3-7. The operation of the circuit is

straight-forward. The ac input voltage e^ produces a current in the input

resistor which is equal to e./R36. Input transistor Q13 draws very little

of this signal current, so the instantaneous output voltage e Q

must assume

the value required to draw this current through the full-wave bridge rectifier

in the feedback path of the amplifier. Thus the current waveform in the

meter is the current in R36 which has undergone full-wave rectification.

The above description of circuit operation is accurate only if the

amplifier has high gain and high input impedance. The current in R36 can

be equal to e VR36 only if the voltage swing at point A is small with respect

to e^. The amplifier gain is adequate if this situation exists. Amplifier

input impedance is large enough when the current drawn by its input terminal

is negligible with respect to the current in R36 . The gain and input

impedance of the three -transistor amplifier used is adequate for driving the

small 100 pA meter. The meter movement itself is the largest cause of

non-linearity.

A few properties of the Operational Rectifier and Meter circuit should

be noted. The dc bias current for Q13 flows through the 0-100 pA meter. This

current is typically 0.3 pA, and is always less than 1 pA. Bias network R39,

R40 (see Figure 3-11 ) allows ripple on the dc supply line to appear at the

base of Q14 . Filtering out this ripple by shunting R4O with a capacitor

would improve the amplifier somewhat, but the improvement is not needed here.

C14 is used to slow down the response of the meter. The meter needle w'ould

otherwise dip annoyingly when the TS0 is supplying the 20 ms interrupted

signal. The meter still responds too quickly, however, to be useful when

the TS0 supplies its 200 ms interrupted signal. Note that the voltage of C14

reverse biases the bridge diodes, requiring the amplifier output voltage to

overcome this reverse bias before signal current can flow through the bridge

network.
1

l

dc blocking

capacitor

-h4-

CI2

current

summing

junction

FIGURE 3-7. Block diagram of Operational Rectifier.
3-17

Power Supply (Q.l6) ». The Power Supply circuit (Figure 3-11) consists of Ql6, C15, R42, and

the battery pack. Ql6 is used as a diode to protect the DU from damage if

the supply voltage is reversed. The diode -connected germanium transistor

has substantially less forward voltage drop than even a high-conductance

germanium diode. C15 maintains low ac impedance on the supply line. R42

provides protection for Q1.6 by limiting the turn-on charging current to C15.

The small-signal resistance of the battery pack can exceed 300 ohms without

affecting the operation of the DU. In other words, as long as the battery

pack voltage tests OK on the built-in meter, the internal resistance of the

cells is not excessive.

M AA” size cells (9/l6" dia x 1-31/32" long) were selected as the power

source because they are commonly available and provide long service life.

These cells are available in several different chemical systems. The table

shown below provides a comparison of several battery systems available.

Carbon-zinc cells designed for use in transistor radios (Eveready #1013 or

equivalent) are recommended for general purpose use. The table can be used

to select cells to meet special requirements, such as need for extra-long

life or high-temperature operation.
Comparison of Battery Systems

Chemical System Eveready Approximate

Operating Life

in Hours*

Suggested

Temperature

Range in °C

carbon-zinc

(transistor

radio)

1015 130 -15 to +40

carbon-zinc

(flashlight

service

)

915 90 -15 to +40

mercury E502

or E9

270 +5 to above +50

alkaline E91 100 -25 to above +50

nickel-cadmium

(rechargeable,

sealed)

C450 45 -20 to +45

Operating life is based on

two hours per day at 2l°C,

a load of 150 ohms across a single cell, cell used

and 1.1 volts at end of life.
TEST DATA

Detector Unit

This section is a collection of data obtained from tests made on an

operating DU. The data were obtained from measurements made on only one DU.

DC Voltage Measurements : The dc voltage measurements are included as a part of the circuit

diagram, Figure 3-11.

AC Voltage Measurements : A series of ac voltage measurements made with an input signal of fixed

frequency and amplitude is given in Figure 3-3.

Frequency Response : Frequency response curves for the DU are given in Figures 3-3, 3-9,

and 3-10.

Noise :With the PROBE connector open-circuited and the GAIN control set to its

extreme clockwise (maximum gain) position, the noise voltage appearing across

a 500 ohm resistive load at the HEADPHONES jack is approximately 18, 11, and

8 mV rms for the 150, 270, and 570 Hz bands respectively. The noise voltage

is around 1.4 mV rms for all band settings when the GAIN control is set to

its extreme counterclockwise (minimum gain) position.
3-20

0 Location mV rms Waveform

1 PROBE connector, pin A 1.00 268 Hz sine wave

2 Tl, pin 5 0.94 268 Hz sine wave

3 Q1 collector .09 268 Hz sine wave, distorted

4 03 emitter 3.6 268 Hz sine wave

5 03 base 5.6 268 Hz sine wave

6 Q3 collector 344 268 Hz sine wave

7 Q,5 emitter 340 268 Hz sine wave

8 R17 sliding contact 2.0 268 Hz sine wave

9 Q6 base .09 268 Hz sine wave

10 Q6 collector 1.16 268 Hz sine wave

11 Q7 base < .02

12 09 base 11.3 268 Hz sine wave, positive peak flattened

13 Q9 collector 830 268 Hz sine wave

14 T2, pin 4 940 268 Hz sine wave

15 Oil base 2.14 268 Hz sine wave

16 Qll emitter 1.83 268 Hz sine wave, negative peak flattened

17 Qll collector 4.3 536 Hz sine wave plus some 268 Hz signal

18 T3, pin 1 1620 268 Hz sine wave

19 T3, pin 4 1480 268 Hz sine wave

20 013 base 3.2 268 Hz sine wave plus large 536 Hz signal

21 015 collector 400 268 Hz rounded square wave, 860 mV p-p

22 016 emitter 3.7 536 Hz sine wave plus some 268 Hz signal

Test Conditions:

1) Battery pack voltage is 6 V dc,

2) Rotary selector switch set at 270 Hz.

3) 500 ohm load connected to the HEADPHONES jack.

4) 1.00 mV rms sine wave applied to the PROBE connector from a 0.5 ohm

source; frequency is adjusted to maximize the panel meter reading.

5) GAIN control adjusted to produce full-scale deflection of the meter.

Notes:

1) Ballantine Laboratories Model 320 true rms voltmeter used to measure

ac voltages.

2) Input impedance of the voltmeter is 10 megohms shunted by approximately

90 pF (includes capacitance of shielded input cable used).

3) All voltages are measured with respect to ground.

4) If an ac operated power supply is used instead of the battery pack,

measurements 3 and 11 will probably be masked by voltage fluctuations

from the supply.

FIGURE 3-8. Typical ac voltage measurements for the Detector Unit.
VOLTAGE

AT

HEADPHONES

JACK

IN

mVrms

3-21

FREQUENCY IN Hz

Test Conditions:

1) A 1 mV rms sine wave test signal is applied to the PROBE

input connector from a 0.5 ohm source.

2) A 500 ohm resistive load is connected at the HEADPHONES jack.

3) The GAIN control is adjusted to produce 1.00 V rms at the

HEADPHONES jack at the response peak of each frequency hand.

FIGURE 3-9. Frequency response of each band of the Detector Unit
VOLTAGE

AT

HEADPHONES

JACK

IN

mVrms

3-22

FREQUENCY IN Hz

Test Conditions:

l) Battery pack voltage is 6.0 V dc.

Rotary selector switch set at 270 Hz.

500 ohm resistive load connected to the HEADPHONES jack.

1.0 mV rms sine wave applied to the PROBE connector from a 0.5 ohm sourc

2 ) 3)

4)

5) GAIN control adjusted to produce 1.00 V' rms at the HEADPHONES jack at the frequency which maximizes the output voltage.

FIGURE 3-10 Frequency response of the Detector Unit near the response peak
1) All switches are mounted on

a common shaft and are mechanically ganged. SI is mounted the closest to the front panel, S3 is farthest from the front panel.

2) Switches are shown in their extreme ccw position as viewed from the front panel.

3) The pointer boxes I—> contain dc voltage measurements, in volts, made at the indicated points under the following test conditions: DETECTOR UNIT

1. Battery pack voltage is 6.00

V dc.

2. PROBE connector is open-circuited (i.e. no ac input signal).

3. Rotary switch is set to 270 Hz.

4. GAIN control is at its extreme ccw (minimum gain) position.

5. Voltages are measured with respect to the circuit common with

a highimpedance voltmeter which does not measurably load the circuit.

3-25

DESCRIPTION OF CIRCUIT COMPONENTS

Detector Unit

Capacitors : Polarized capacitors are solid tantalum electrolytics from the Sprague

150D series (military series CS13). Voltage ratings are 35, 20, and 10

volts dc for the 1, 10, and 100 ^lF capacitors, respectively.

Cl, C2, and C3 are 50 volt dc polycarbonate capacitors whose capacitance

at -25 and + 85°C is within 0.5$ of their +25 °C value. Cl and C2 are

Component Research Company, Inc. part number 05PL224D, and C3 is part

number 05PL104D.

C4 and C9 are miniature tubular capacitors from the Sprague 192P "Pacer"

series and have ±10$ tolerance and an SO volt dc working voltage.

C13 is a disk ceramic capacitor.

Resistors : Unless otherwise noted, 0.25 watt carbon composition resistors having

±5$ tolerance are used.

Resistors which are specified by the circuit diagram to have 1$

tolerance are Corning Electronic Components type NC4. These are 0.1

watt tin oxide resistors having temperature coefficients no greater

than ±50 ppm/°C.

R17 is a 2 watt molded carbon potentiometer having a clockwise

logarithmic taper, "0" ring seals on shaft and bushing; AlienBrad ley JA1P056P103AA.

Transformers : T1 is a Stancor PCT-54 miniature transistor transformer. This transformer

has a 12 ohm winding (2.7 ohms dc and 8 mH) and a 600 ohm center

tapped winding (92 ohms dc).

T2 is a Triad SP-13 miniature transistor transformer. This transformer

has an 800 ohm winding and a 20,000 ohm center-tapped winding.

T3 is a Stancor PCT-62 or a Triad SP-52 miniature transistor transformer.

This transformer has a 600 ohm winding (97 ohms dc) and a 1500 ohm

center-tapped winding (167 ohms dc).
3-26

Miscellaneous : F, the low-pass rf suppression filter is Erie Technological Products,

Inc. part number 1200-019. This is an L-section attenuator made

up of a capacitor of- around 2.2pF and an inductor of around 80pH

and 1.8 ohms dc.

The meter is a Honeywell model HS1, 0-100 pA meter. This is a 1.5"

unit which is ruggedized and hermetically sealed.

The PROBE connector, Jl, is military type MS-3102A-10SL-4P (Amphenol,

Cannon, etc.

)

The three-deck rotary switch, Sla through S3b , is assembled from the

following Centralab components: Two PS-22 switch sections, one PS-23

switch section, and one P-504 shaft and index assembly. The PS-23

switch section is mounted closest to the knob end of the shaft. All

of the switch sections are 2-pole, 5-position units. The PS-22 switch

has shorting (make-be fore -break) contacts, while the PS-23 switch

has non-shorting (break-before-make) contacts.
Section 4

MISCELLANEOUS
4-2

IMPROVEMENTS SUGGESTED for the CABLE FAULT LOCATOR

Test Signal Oscillator :

l) Increasing the secondary voltage of power transformer T1 to 20 V rms

each side of the center tap when the primary voltage is 130 V rms

will enable the instrument to operate on input voltages between 100

and 130 V rms, and also increase the output power available from the

TS0 by around 20$.

Detector Unit :

1) Connecting C4 between the Q3 collector and ground will bring the

570 Hz band much closer to its stated center frequency.

2) The noise generated by the Tuned Input Amplifier can be reduced.

Transistor Q1 is the predominant noise source and the use of a lower noise transistor is desirable. Employing a lower noise

transistor for Q3 causes a small additional reduction in the

amplifier noise. For example, using 2N43&4 transistors for Q1

and Q3 results in typical noise voltages of 6, 5, and 3.5 mV rms

for the 150, 270, and 570 Hz bands respectively. This compares

with typical noise voltages of 18, 11, and 8 mV rms when 2N930

transistors are used. The preceding noise voltages are measured

across a 500 ohm resistive load at the HEADPHONES jack when the

GAIN control is at its extreme clockwise (maximum gain) position

and the PROBE connector is open-circuited.
4-3

1. REFERENCES

Guide to Use of the AN/TSM-11 Cable Test-Detecting Set . James E. David

and J. W. Simeroth, National Bureau qf Standards Report 85 96, December

1964.

2

Radiotron Designer’s Handbook , fourth edition, edited by F. LangfordSmith, reproduced and distributed by Electron Tube Division of Radio

Corporation of America, Harrison, N. J. , pages 620 - 621.

3.

A Quick Review of Parallel-T Filters . Dean Mensa, EDN Magazine,

March 1968, pages 112 - 114.

4.

Electronic Designer’s Handbook . Robert W. Landee, Donovan C. Davis,

Albert P. Albrecht, McGraw-Hill Book Company, Inc., 1957, pages 16-23,

16-24, 16-26, 16-27, 16-28.

5.

Vacuum Tube Amplifiers . George E. Valley, Jr. and Henry Wall man, McGrawHill Book Company, Inc., 1946 pages 364 - 397.

6.

"Eveready" Battery Applications and Engineering Data . Union Carbide

Corporation, 1965.
.

'