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THE BRADLEY DEPARTMENT

OF ELECTRICAL ENGINEERING

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STUDY OF SPREAD SPECTRUM MULTIPLE ACCESS

SYSTEMS FOR SATELLITE COMMUNICATIONS

WITH OVERLAY ON CURRENT SERVICES

by Tri T. Ha and Timothy Pratt

VIRGINIA POLYTECHNIC INSTITUTE AND STATE UNIVERSITY

BLACKSBURG, VA 24061

prepared for

NATIONAL AERONAUTICS AND SPACE ADMINISTRATION

NASA Lewis Research Center

"E_i:!

_!ii_i Contract NAS 3-24887

_ULTIFLE ACCESS SYSTE_ FC_ 5_£LITE

CC.SMU_C_It_S- _I_H:. OV_H/_AY. _1_ CU_R.E_T

.c_VlC_.C {Virginia _clytechnic Inst. and

_ta.t¢ Univ.) 255._

_89-23_57

laclas

CSCL 17B G3/32 02115_8

VIRGINIA POLYTECHNIC INSTITUTE AND STATE UNIVERSITY

Blacksburg, Virginia 24061 (703)961-6646
Final Report

1. Report No_ 2. Government Accession NO.

CR 180-827

4. Title and Subtitle

Final Report on Contract NAS3-24887: "A Study of Spread

Spectrum Multiple Access Systems for Satellite

Communications with Overlay on Current Service."

7. Author(s)

Tri T. Ha, Timothy Pratt

9. Performing Organization Name and Address

Virginia Polytechnic Institute and State University,

Blacksburg, VA 24061

12. Sponsoring Agency Name and Address

NASA Lewis Research Center

Cleveland, OHIO 44135

I 3. Recipient's Catalog No.

5. Repo_ Date

December 1987

_6. Performing Orgamzation Code

8. Performing Organization Report No.

10. Work Unit No,

11. Contract or Grant No.

NAS3-24887

13. Type of Report and Period Covered

Contractor Report

14. Sponsoring Agency Code

15. Supplementa_ Notes

Project Manager: John E. Zuzek, NASA Lewis Research Center,

21000 Brookpark Rd.; Cleveland, OH 44135

16. Abstract

This report investigates the feasibility of using spread spectrum techniques

to provide a low-cost multiple access system for a very large number of low data

terminals.

Part A of this report presents two applications of spread spectrum technology

to very small aperture terminal (VSAT) satellite communication networks. It

describes two spread spectrum multiple access systems which use a form of

noncoherent M-ary FSK (MFSK) as the primary modulation and analyzes their

throughput. The analysis considers such factors as satellite power constraints and

adjacent satellite interference. It considers the e_fect of on-board processing on

the multiple access efficiency and investigates the feasibility of overlaying low

data rate spread spectrum signals on existing satellite traffic as a form of

frequency reuse.

Part B of the report examines the use df chirp for spread spectrum

communications. In a chirp communication system, each data bit is converted into

one or more up or down sweeps of frequency, which spread the RF energy across a

broad range o_ frequencies. Several different forms of chirp communication

systems are considered, and a multiple-chirp coded system is proposed for overlay

service. The mutual interference problem is examined in some detail and a

performance analysis undertaken for the case of a chirp data channel overlaid on

a video channel.

17. Key Words (Suggested by Authors)) 18. Distribution Statement

Spread Spectrum Multiple Access Satellite

Communications Overlay Service

19. SecurityUnclClassif. assi (offiedthis repOrt) 120. Security Classif. Uncl assi (of this fiedpage) 121. No. of236pages

"For sale by the National Technical Information Service, Springlield, Virginia 22161

Abstract

This report investigates the feasibility of using spread spectrum techniques to provide a low-cost

multiple access system for a very large number of low data terminals.

Part A of this report presents two applications of spread spectrum technology to very small aperture

terminal (VSAT) satellite communication networks. It describes two spread spectrum multiple access

systems which use a form of noncoherent M-ary FSK (MFSK) as the primary modulation and analyzes

their throughput. The analysis considers such factors as satellite power constraints and adjacent

satellite interference. It considers the effect of on-board processing on the multiple access efficiency

and investigates the feasibility of overlaying low data rate spread spectrum signals on existing satellite traffic as a form of frequency reuse.

Part B of the report examines the use of chirp for spread spectrum communications. In a chirp

communication system, each data bit is converted into one or more up or down sweeps of frequency,

which spread the RF energy across a broad range of frequencies. Several different forms of chirp

communication systems are considered, and a multiple-chirp coded system is proposed for overlay

service. The mutual interference proi_tem is examined in some detail and a performance analysis

undertaken for the case of a chirp data channel overlaid on a video channel.

Abstract
Table of Contents

Part A. SPREAD SPECTRUM MULTIPLE ACCESS AND

OVERLAY SERVICE

I. Introduction

Spread Spectrum Multiple Access

Overlay Service

PAGE

Z

3

4

II. Overview of Multiple Access and Spread Spectrum Techniques

Conventional Multiple Access Techniques

On-Board Processing

Overlay Systems

5

5

11

12

III.MFSK Spread Spectrum Multiple Access

Preliminaries

Performance Analysis

13

13

16

Table of Contents

Ill
Multiple Access Systems

System 1: MFSK/DS/FDM

System 2: MFSK/DS/FH

Numerical Examples

Example 1: C Band System

Example 2: Ku Band System

Sample Link Analyses

C-band System-968 users at 1.2kbps

Ku-band System-57 users at 56 kbps

Conclusions

22

22

23

26

27

27

32

34

35

37

IV. Spread Spectrum Overlay Service

Overview

Comparison of Spread Spectrum Overlay Systems

Frequency Hopped Spread Spectrum

MFSK/DS Spread Spectrum Overlay

Direct Sequence Overlay

Interference Modeling

Modeling of Wideband Signals Passed Through Narrowband Filters

Modeling of interference PSK Signals by Multiple PSK Interferers

Overlay System Performance Analysis

Overview

Description of Existing Signals

Overlay on FDM/FM/FDMA Telephone Traffic

FDM Overlay System: C-Band

Overlay of Digital SCPC System

SCPC Overlay System: C-band

Analog Television Overlay

38

.38

39

39

40

41

41

43

"43

44

44

44

45

51

52

56

57

Tabla of Contents I v
Analog Television Overlay: C-band

Extension to Multiple Overlay Signals

Conclusions

61

62

62

V. On Board Processing for Multiple Access Systems

Introduction

Simple Procession Transponder

Dual Modulation Systems

MFSK/TDM Processing Transponder System

Ku Band MFSK/DS-TDM System: 110 user_

64

64

68

68

71

72

Vl. VSAT Networlr_An Overview

Introduction

Architectures

Ku-Band versus C-Band

Power-Limited Networks

Spread Spectrum VSAT Networks

Epilogue

74

74

75

76

81

86

92

VII.Conclusions

-94

IIX.References

96

Appendix A. Performance of Noncohe_mt MFSK In Colored Noise 99

Appendix B. Derivation of Interference PSD

103

Appendix C. MFSKIDS/FOM Performance _ Program 108

Table of Contents v
Appendix O. MFSK/DS/FH Performance Analysis Program

IIi

Table of Contents

vi
Part B. Chirp Spread Spectrum System and Overlay

Service

1. Introduction

PAGE

117

2. Review of Chirp Spread Spectrum Systems

120

3, Coded Multiple Chirp Spread Spectrum System

3.1. Basic System Concept

3.2. Performance Analysis

3.2.1. System Parameters and Waveform Definitions

3.2.2. Cross-correlation of Up Chirp and Down Chirp

3.2.3. Cross-correlation between Code(Word) Sequence

3.2.4. Chip Error Probabili_(Dechirper BER)

3.2.5. Performance of Digital Code Correlator

3.2.5. Chirp Spread Spectrum Multiple Access

127

127

131

i31

132

136

137

142

146

4. Oveday Service-An Overview

148

S. Interference Analysis I ( _ Approach )

5.1. Analog FM-TV Demodulat)on

5.2. Chirp Interference to Analog ;:M.1V Signal

5.3. Simulation of FM-Discnmmator Output

5.4. Analog FM-TV Signal Lnterference on Chirp Demodulation

150

151

151

162

169

Table of Contents v l I
6. Interference Analysis II ( Modeling and Simulation )

6.1. Analog FM-TV Signal Modeling I - Without Preemphasis

6.2. Simulation of Wideband FM-TV Signal and Chirp I - No Preemphasis

6.3. Analog FMoTV Signal Modeling II -With Preemphasis

6.3.1. Preemphasis Characteristic

6.3.2. Modeling of FM-TV Video Signal

6.4. Wideband FM-TV Signal Simulation II - With Preemphasis

6.5. Overlay System Link Analysis

6.5.1. FM-TV Overlay Link Analysis

6.5.2. Chirp S.S. Overlay Link Analysis

174

175

178

189

189

195

20O

203

204

206

7. Performance Analysis of Overlay Service I - Without Preemphasis

7.1. Signal Power Calculation

7.2. Interference Power Calculation

7.2.1. Chirp Interference to FM-TV

7.2.2. FM-TV Interference to Chirp

209

210

213

213

214

8. Performance Analysis of Oveday Service II - With Preemphasis

8.1. Signal Power

8.2. Interference of Chirp to FM-TV

8.3. Interference of FM-TV to Chirp

222

222

225

225

9. Conclusions 232

10. References

233

Table of Contents I I x
List of Illustrations -- Part A

Fig. 1. FDMA Signal Organization[2]

PAGE

6

Rg. 2.

Rg. 3.

Typical TOMA System[5]

SSMA System Model

8

15

Rg. 4, SSMA System ModeI-N users

17

Rg. S.

Rg. 6.

MFSK Receiver Model

Interference PSD Model

18

21

Rg. 7. MFSK Slot Bandwidth

24

Fig. 8. Hybrid MFSK]DS/FH SSMA System 25

Fig. 9 Performance of MFSK/DS/FDM System(C band) 28

Fig. 10. Performance of MFSK/DS/FH Systems(C band)

29

Listof Illustrations-- Part A ix
Fig. 11. Performance of MFSK/DS/FDM Systems(Ku band) 3O

Fig. 12, Performance of MFSK/DS/FH Systems(Ku band)

31

Fig. 13.

Fig. 14.

Full Mesh Network Architecture

DS Overlay System

33

42

Fig. lS. FDMA Existing Signal

46

Fig. 18. DS Overlay Signal 47

Fig. 17. Interference PSD caused by the FDMA Signal

49

Fig, 18. SCPC Existing Signal

53

Fig. lg, Interference PSD caused by the SCPC Signal

55

Fig. 20. BPSK Signal Passed Through an FM Demodulator

59

Fig. 21. Interference PSD Caused by a "IV Signal 60

Fig. 22. Classical Transponder

65

Fig, 23.

Fig. 24.

Processing Transponder

Multipoint to Point Network Architecture

66

7O

Fig. 25. Single Hop Architecture

77

Fig, 26. Hybrid Architecture 78

List of Illustrations -- Part A x
Fig.27. FullMeshArchitecture 79

Fig. 28. SSMA Modem 87

List of Illustrations -- Psrt A x i
List of lllustrations -- Part B

RI. 1. Within Chirp Modulations

PAGE

123

Flg. 2a. Multiple Chirp Modulations i24

Fig, 2b, Typical Waveforms of Multiple Chirp Modulations

125

Rg, 3a, Coded Multiple Chirp S S. System 129

Rg, 3b, Typical Waveforms of Coded Multiple Chirp S.S. System (proposed)

130

Fig, 4, Autocorrelation (matched case) and Crosscorrelation (mismatched case)

135

Fig, 5a. Chip Error Probability of Dechirper 140

Fig, 5lb, BER Comparison of Noncoherent FSK and Chirp for different BT product 141

Fig. 6. Word Miss Probability vs. DecnirperBER

144

Fig. 7, Word False Alarm Probamlity vs Miss Probability 145

List of Illustrations-- Part B x I I
Fig. 8. Time waveform of FM-discriminator output error voltage for unmodulated TV carrier plus

chirp interference 153

Fig. 9. Time waveform of FM-discriminator output error voltage for FM-TV carrier plus chirp

interference ( re(t)= 0.5V dc ) 154

Rg. 10. Simulated 9 step staircase function (time plot) 164

Fig. 11. Received and Iowpass filtered simulated 9 step staircase function 165

Flg. 12. Error output voltage of FM-discriminator after bandpass filtering 166

RO. 13. Spectrum of error output after bandpass filtering (3-4 MHz) 167

Fig. 14. Spectrum of 9 step staircase function 168

Rg. lie Typical compression filter output for chirp signal input 172

Fig. 111. Compression filter output for FM-TV signal with amplitude = 0.1V 173

Rg. 17. Time plot of simulated 200 step staircase function model for video signal 177

RO. 18. Time plot of simulated 256 step random amplitude model for video signal 182

Rg. 19. Transmitted FM-TV signal time plot (random) 183

Fig. 20, Frequency spectrum of Fig. 19 (random) 184

Fig. 21. Transmitted FM-TV signal time plot (linear) 185

Fig. 22. Frequency spectrum of Fig. 21 (linear) 186

LiM of Illuma'atlon= -- Part B x i I v
Fig. 23.

Fig. 24.

Transmitted chirp signal time plot

Frequency spectrum of Fig. 23

187

188

Fig. 25. Typical NTSC modem structure with preemphasis/deemphasis

191

Fig. 26.

Rg. 27.

TV preemphasis characteristics (525, 625, B19 and 405 lines)

NTSC 525 line - CCIR REC.405 preemphasis characteristics

192

193

Fig. 28.

Rg. 29.

Simulated preemphasis filter for NTSC 525 line TV system

TV message m(t) before preemphasis (time plot); random amplitude

194

196

Fig. 30. "IV message m(t) after preemphasis (time plot); random amplitude

197

Fig. 31.

Rg 32.

TV message m(t) before preemphasis (time plot); linear staircase fn

TV message m(t) after preemphasis (time plot); linear staircase fn

198

199

Fig. 33.

Fig. 34.

Transmitted FM-TV spectrum (re(t) = random, preemphasis version)

Transmitted FM-TV spectrum (re(t) = linear, preemphasis version)

201

o.

202

Fig. 3S.

Fig. 36.

36dB attenuated chirp s_gnal spectrum

Overlapped spectrum of FM-TV (staircase re(t)) and 36dB atten chirp

212

217

Fig. 37. Overlapped spectrum of FM-T'V (random amplitude m(t))and 36dB arten chirp

218

Fig. 38. C/I ratio vs IF offset frequency (random amplitude message and linear staircase) 221

Listof Illustrations-- Part B x iv
Fig.39. Overlappedspectrumof FM-TV (preemphasis) and 36dB atten chirp (random)

Fig. 40.

223

Overlapped spectrum of FM-TV (preemphasis and 36dB atten chirp (linear staircase) 224

Fig, 41. C/I ratio vs IF offset frequency (preemphasis case) 229

Fig, 42. C/I ratio vs IF offset frequency (preemphasis case and no preemphasis case) 230

Fig. 43. Offset (IF) frequency vs chip error rate for some compression gain 231

List of Illustrations.- Part B

XV
Summary

The major objective of this study is to investigate the feasibility of using spread spectrum techniques

to provide a low-cost multiple access system for a very large number of low data rate terminals. In

addition, the study also examines the feasibility of overlaying spread spectrum transmissions onto

existing communications and the use of on-board processing to improve the system throughput. The

• report considers three spread spectrum techniques; 1) MFSK]DS, 2) hybrid MFSK]DS-FH, 3)Chirp. Part

A of the report addresses mainly the first two techniques and Part B addresses the third technique.

The MFSK/DS and hybrid MFSK/DS-FH systems proposed can achieve a throughput of

approximately 6-8% for Ku-band satellites and 2-3% for C-band satellites. The advantage of spread

spectrum systems is that they make very small aperture terminal(VSAT) networks possible with

antenna diameter less than 1.8 meters for C-band operation and when the angular satellite spacing

is small, situations which makes VSAT networks infeasible for other modulations. At Ku-band, spread

spectrum systems may offer a cost sawng advantage over single-channel-per-carrier (SCPC) systems

in a full mesh architecture. This is poss_bte because an MFSK/DS or a PSK/DS VSAT does not require

a frequency synthesizer as an SCPC VSAT does. This advantage is lost in a star architecture operating

at Ku-band. The principal conctus=on reached here is that VSAT networks are inherently power-limited

and not bandwidth-limited, therefore Jt makes Iittte difference whether non-spread spectrum or spread

Summary x v i
ORIGINALPAGEiS

OF POORQUALITY

spectrumtechnology is used at Ku-band. At C-band, the use of spread spectrum technology for VSAT

networks is almost unavoidable for antenna diameters less than 1.8 meters.

The use of PSK/DS overlay signals on existing traffic is also feasible. When convolutional coding

is used, a data rate of 10 to 20 kbps for an overlay signal may be achieved per satellite transponder,

On-board processing has the ability to dramatically increase the throughput of spread spectrum

systems ( almost twice as much ). But the future of on-board processing in the commercial market,

however, is probably not bright because the complex on-board electronics required would make the

satellite very expensive and less reliable. Furthermore, the transponder becomes dedicated to a

certain type of service and hence less flexible in handling different kinds of traffic.

Chirp communication systems use swept frequency techniques to spread the RF energy across a

much wider bandwidth than that occupied by the data signal. Each data bit is converted to one or more

chirps, and each chirp is a linearly increasing or decreasing sweep across a predetermined RF band

of frequencies. Thus all chirp systems are inherently spread spectrum. The chief advantage which

chirp offers over other spread spectrum techniques Is passive synchronization. All direct-sequence

spread spectrum system must first synchronize the receiver to the transmitted sequence before any

data can be extracted. In a chirp system, the despreading operation is achieved with a

pulse-compression filter, which is a pass,re device requiring no synchronization at its input. This is

a major advantage in a system which employ1 many transmitters and receivers with low duty cycles

and low data rates.

A simple chirp spread spectrum offers good compatibility with other modulated signals; however,

for multiple access to a common channel orthooonal between different chirp signals is needed. This

report examinee several ways in which such orthogonality can be provided, by using within-chirp and

multiple.chirp coding. Multiple-coded ch,r!_ was selected as a candidate system for a low bit rate

overlay service, and its performance was studied to derive the parameters of working system.

Multiple-coded chirp can be used successfully for spread-spectrum overlay on an existing channel.

This report examines in detail the mutual ,nterference resulting from overlaying a multiple-coded chirp

signal on an occupied FM-TV satell,te ct_annol The performance, in terms of interference ratios for the

xvil

Summary
TVsignalandBER for the overlay data channel, has been derived for a variety of bit rates and C/I

ratios.

Finally, a typical application is presented in which data from a VSAT earth station is sent back to a

large earth station over a TV channel using a C-band satellite transponder. It is shown that a data rate

of 2.4 kbps can be achieved with a low power transmitter(300mW), with no significant interference at

adjacent satellites.

Summaw xl x
Part A. Spread Spectrum Multiple Access and Overlay

Service

Psrt A. Spresd Spectrum Multiple Access and O_erlay Service
I. Introduction

Spread spectrum signals have been used in military satellite communications for years because of

their ability to reject interference and jamming. In commercial satellite systems, where there is no

intentional jamming, spread spectrum techniques were thought to provide no advantage.

For high-capacity point to point communications, satellite and fiber optic communications are cost

efficient, since the substantial cost of several large earth stations or the installation of miles of optical

fiber can be divided among many users. In large "thin route" networks, where the terminals are

widely separated-and the data rates are relatively low, the network overhead cost becomes

prohibitive. The geographical spacing increases installation and networking costs for an optical fiber

system dramatically. In satellite communication, the main problem would be the large number of

users. The high cost of many large earth stations could not be justified for terminals transmitting low

rate data.

Recently there has been a great deal of =nterest in satellite communication from very small aperture

terminals. This paper analyzes the performance of two types of satellite communication from such

terminals using spread spectrum tect_n_ques and shows that spread spectrum provides some unique

advantages that make these services feasible In addition, a small aperture terminal network using

a processing transponder is presented and analyzed.

I. Introduction 2
Spread Spectrum Multiple Access

The majority of current satellite traffic consists of analog television or medium to high data rate

signals transmitted from a small number of large aperture earth stations. Mainly because of their

large antenna diameter, these earth stations are very expensive. When there are many users, the

cost of this type of earth station would make the network prohibitively expensive. Since the earth

station cost is most dependent on the antenna diameter, the obvious way to reduce its cost is to

reduce the antenna diameter. Earth stations that use small diameter antennas have become known

as "Very Small Aperture Terminals', or VSATs.

Reducing the antenna diameter affects the system performance in two ways. First the antenna gain

is reduced, lowering the link carrier to noise ratio. Second, the antenna beamwidth is broadened,

increasing both the interference received from and transmitted to adjacent satellites. The signal

transmitted from a small aperture earth station interferes with the operation of adjacent satellites

more than the signal from a large aperture terminal on the uplink. Signals transmitted from adjacent

satellites will not be attenuated significantly, increasing the interference level in the VSAT receiver.

The interference rejection capabilities of spread spectrum signals make multiple access from small

earth terminals feasible for a large number of low data rate users. Chapter 3 analyzes the

performance of two spread spectrum multiple access systems which use a form of noncoherent M.ary

FSK as their primary modulation. These performance results are then applied to practical example

satellite systems using small diameter antennas.

I. Introduction 3
Overlay Service

Overlay service is a method of frequency reuse which involves adding a signal to a transponder

already considered to be full by overlaying the new signal within the bandwidth of the existing signal.

For this type of service to work, the overlay signal must not interfere appreciably with the existing

signal, and vice-versa. A spread spectrum signal with a wide, fiat power spectral density (PSD) can

appear noiselike to the existing signal, and will not disturb the existing signal if the power is low

enough. Since the overlay signal power will be much lower than that of the existing signal, the overlay

signal must be able to reject the strong interference caused by the existing signal. The VSAT receiver

can use the processing gain of spread spectrum systems to achieve the required interference

rejection. Chapter 4 compares various types of spread spectrum signals to find the type most suitable

for overlay systems. It finds the maximum overlay data rate for three types of existing signals and

calculates the improvement in overlay data rate achieved by using a representative convolutional

code.

h Introduction
II. Overview of Multiple Access and Spread Spectrum

Techniques

Conventional Multiple Access Techniques

The two major multiple access techniques in commercial use today are 1) frequency division multiple

access, and 2) time division multiple access. Frequency division multiple access (FDMA) is the most

mature of the multiple access technologies. In this system, each user is allocated a specific part of

the transponder bandwidth for its exclusive use [1]. Figure 1 illustrates a typical four channel system.

FDMA systems require a low level of coordination among users, with only frequency band

assignments to be made. The transmitting and receiving equipment is essentially the same as when

there is no multiple access.

The major disadvantage of FDMA systems is that the transponder output power must be reduced, or

"backed-off', to ensure that the transponder is operating in its linear region. If the transponder is

sufficiently nonlinear, intermodulation interference between different users will result [3].

II. Overview of Multiple Acceu end Spread Spectrum Techniques 5
t I

FM FM

8CD | _" 1

A

t

I=1

1=2

1=OMA

BASE BANDS

I I

FM FM

:ol:ol

t t 4"" CARRIERS"°""

IW3 F4

Figure 1. FDMA Signal Organization [2]

II. Overview of Multiple Access and Spre_l Si)ecuum Techniques 8
Time division multiple access (TDMA) avoids the backoff problem inherent in FDMA systems by

allowing only one user to occupy the transponder at a time, so that no interference between users is

possible. Each user is allowed to use the entire transponder bandwidth in sequence for a short period

of time. This requires accurate time synchronization between all users. Synchronization

requirements make the earth terminals expensive and complicated [4]. Figure 2 illustrates the time

multiplexing of TDMA signals transmitted from a three channel system.

As mentioned previously, to construct a network of small aperture earth terminals, which have wide

antenna beamwidths, a modulation type which reduces adjacent satellite interference is needed. The

FDMA and TDMA systems described above use conventional modulation types, such as FM for analog

signals, and QPSK for digital signals. These systems use these modulations for reasons other than

interference rejection. Interference control is achieved primarily by reducing the antenna gain in the

direction of the adjacent satellites, several degrees off axis, by using very large diameter antennas.

The interference transmitted to adjacent satellites in VSAT networks can be reduced if the PSD of the

transmitted signals is relatively low and fiat.

Spread spectrum signals have features which can effectively reduce these problems. Some spread

spectrum signals have relatively fiat PSDs. In addition the correlation of the received signals with

spreading codes in spread spectrum receivers effectively distributes the energy of uncorrelated

interfering signals over a large bandwidth whtle simultaneously compressing the bandwidth of the

desired signal. This "processing gain" =s a slgn=ficant source of interference rejection. In addition,

most spread spectrum multiple access systems have little network overhead. Transmitters can

operate asynchronously, unlike TDMA systems, and do not need to coordinate the operating

frequencies as in FDMA systems.

Any type of spread spectrum signal can De used in a spread spectrum multiple access system. The

two major classifications of spread spectrum s=gnals are direct sequence and frequency hopped.

Direct sequence systems spread a s_gnal s 13andwldth by mixing the signal with a relatively high rate

II, Overview of Multiple Access and $1:_ed Sl_ctrum Techniques 7
Figure 2. Typical TDMA System [5]

II. Overview of Multiple Access and Spread Spectrum Techniques S
pseudonoise (PN) code [6]. Frequency hopped systems spread the signal by using a carrier signal

which hops around the channel bandwidth in a pseudorandom manner [7].

In satellite spread spectrum multiple access, also known as code division multiple access (CDMA),

uplink earth stations transmit signals that share the same bandwidth and time. Each transmitter

spreads its signal with a unique code. Differentiation between desired and undesired signals is

accomplished at the receiver, which correlates the received signal with a synchronized version of the

spreading code. The bandwidth of the correlated signal is collapsed to its unspread bandwidth, while

the bandwidth of the other users' signals is spread further, decreasing the PSD of the interference [8].

Direct sequence SSMA systems can be either "sequence synchronous" or "sequence asynchronous"

[9]. Sequence synchronous systems require the bit transitions of each transmiter's spreading code

to be in synchronization when received by the satellite. These systems can achieve very low levels

of inter-user interference if orthogonat sl_reading codes are used, but this limits the number of users

to the length of the code [10]. Synchronous systems are impractical for large thin-route networks

however. The synchronization of hundreds to thousands of widely spaced transmitters is an

overwhelming problem, especially when the earth station cost is a major consideration. Loss of

synchronization produces a catastrogh=c _ncrease in the users' error rate [11].

Sequence asynchronous DS SSMA systems allow completely asynchronous transmissions. Inter-user

interference is increased significantly over orthogonal synchronous DS SSMA however. The amount

of inter-user interference is dependent on the partial cross-correlations between the different

spreading codes [12,13]. Sequence asyncl_ronous OS SSMA systems are promising for practical

SSMA small earth terminal systems.

Frequency hopped multiple access _I:HMA) =s commonly used with M-ary FSK as the primary

modulation. The available channel t_andw_attt ,s divided into "slots". The MFSK signal hops from slot

to slot under the control of its un=que ;_N co_e Symbol errors occur when more than one user hops

into the same slot. Slow hop FHMA systems hop in frequency no more than once per symbol. Fast

II. Overview of Multiple Access =nd $¢¢oe/ SCqN=:_mtTechniques 9
hop FHMA systems hop more than once per information symbol, and the receiver uses a majority

logic decision circuit to estimate the transmitted symbol based on the multiple received hops. [14]

While MFSK FHMA receivers can be simplified by using noncoherent detection, the cost of a frequency

synthesizer that can produce carrier frequencies over the entire 36 or 54 MHz bandwidth of a satellite

transponder may be expensive, and not suitable for large thin-route networks.

One type of sequence asynchronous DS SSMA system that shows promise for multiple access from

small earth terminals is an MFSKJDS SSMA system. In this system an MFSK information signal is

spread by a PN code. This is the type of system analyzed in Chapter 3.

The performance of individual MFSK/DS signals in the presence of various types of jamming signals

is well known [15]. The performance of MFSK/DS multiple access systems, however, has not been

thoroughly studied. At the time this research was started, the only work on MFSK/DS spread

spectrum multiple access systems is a paper by Yamauchi et al. translated from Japanese [18]. This

paper shows that the multiple access efficiency of MFSK/DS SSMA systems improves when the

spacing between MFBK symbol frequencies is greater than the symbol rate. The paper presents an

analysis of the performance of an MFSK/DS SSMA system which references several sources available

only in Japanese. Because of this, their analysis could not be easily verified.

The analysis presented in Chapter 3 is based on the paper by Yamauchi. The MFSK system with

modified symbol frequency spacing that he presents, which is called *wideband MFSK* here, is used.

The performance equations are completely rederived, with the mathematical details presented in

appendices A and B, and do not agree with those in Yamauchi's paper. All the formulas used in this

report are derived herein.

In addition, the MFSK/DB SSMA system results are applied to satellite multiple access from small

earth terminals. Two original SSMA systems based on the wideband MFSK/DS SSMA system are

II. Overview of Multiple Access and Spread Spectrum Techniquel 10
presentedandanalyzed.Thisanalysisconsidersthepowerandinterferenceconstraintsof practical

satellite systems.

On.Board Processing

The practice of demodulating the uplink signals at the satellite and retransmitting them is known as

•on-board processing'. Simply demodulating the uplink signal and retransmitting it with the same

modulation can give about a 3 dB improvement in C,'N when the uplink C/N and downlink C/N are

nearly equal.

Satellite Switched TDMA (SS-TDMA) is a technique that directs each TDMA burst to its particular

destination through a narrow, high gain spot beam. The high gain downlink antenna significantly

improves the downlink C/N. Since the downlink antenna no longer provides coverage of the entire

service area, SS.TDMA requires the satellite to recognize the intended recipient of each uplink burst

and direct it to theappropriate spot beam [17].

A processing transponder for multiple access of low data rate mobile users has been proposed [18].

This technique requires the transponder to compute a discrete Fourier transform of its bandwidth to

demodulate MFSK uplink signals• Such a technique is said to be much more able to track the doppler

shift of the mobile users' signals.

The analysis of satellite links that use processing transponders is easily accomplished by considering

the uplink and downlink bit error rates separately [19]. Chapter 5 discusses various on-board

processing systems which may improve the throughput of the systems analyzed in Chapter 3 and the

conditions under which they may be used. An analysis of the Ku band example from Chapter 3 is

It. Overview of Multiple Access and Spre_¢l Spectrum Techniques

11
performed to show the possible throughput improvement provided by a i_articular type of processing

transponder.

Overlay Systems

Satellite signal overlay has not been widely studied or practiced. The only type of overlay signal

known to be in use is in a system which overlays a DS signal on TDMA traffic for TDMA loop-back

synchronization [20].

No analytic work on satellite overlay systems was found. One brief reference to a satellite television

overlay experiment gives so few details of the system parameters that it is of no use [21]. The overlay

analysis in Chapter 4 is intended to give a rough estimate of the performance of satellite signal

overlay from small earth terminals. The possible overlay system performance of several spread

spectrum overlay signals is compared. A general methodology for analyzing the overlay performance

is presented and applied to three example systems. The methodology and analysis are original.

II. Overview of Multiple Access and $_r_e4 Sl_u_J,m Techniques 12
I!1. MFSK Spread Spectrum Multiple Access

Many spread spectrum multiple access (SSMA) systems have been studied. Unfortunately the

efficiency, measured by the number of users accomodated in a given bandwidth, is usually low. This

chapter analyzes two SSMA techniques which use a form of noncoherent M-ary FSK as the primary

modulation to improve throughput. It includes a systems level description of the performance of these

systems, while mathematical details are kept to a minimum. The following sections describe and

analyze two practical $SMA systems. The analysis includes calculations of their bandwidth efficiency

and link analyses to assess power and adjacent satellite interference limitations which may occur in

practical systems.

Preliminaries

Several features are essential to keep the system as simple as possible and to reduce the equipment

cost to a level appropriate to small earth term=nals. First, noncoherent detection of the MFSK signal

is assumed, simplifying the receiver Atso, all users transmit asynchronously, with no network

synchronization. Synchronization of hundreds to thousands of transmitters, even at the millisecond

lU. MFSK Spread Spectrum Multiple Acceu 13
level,wouldbeextremelydifficult. Theanalysisassumesthatall usershave the same power at the

earth station receiver input. This assumption is valid for satellite communications since some form

of power control at the transmitters is necessary to prevent power-hogging in the transponder.

Both of the SSMA systems to be analyzed are based on the system model shown in Figure 3.

Information at a bit rate Ra is modulated into MFSK symbols at a rate

R,, where R, = Ra / k and k = log, M. The spectrum of this MFSK signal is spread when mixed with

a PN code at a code rate R=. The j_' user's signal can be described by

oo

sl (t) = "J_ _ rect(t - k_) cos( 2_rfl t ) PN] (t)

ks1

(3.1)

where PNj (t) is a full period of a bipolar NRZ rectangular pseudonoise pulse sequence at a rate R=

, C is the signal power, and 7", = 1/R,.

In a conventional MFSK system the signal is represented by M tones separated in frequency by R,.

This frequency spacing minimizes the total signal bandwidth while maintaing the orthogonality

between the symbols. Minimizing the signal bandwidth is of no concern here since we intend to

spread the signal bandwidth anyway. The modulation can be generalized to allow frequency spacing

between MFSK symbols of

_f = pR s (3.2)

where p is an integer, pal corresponds to the conventional narrowband MFSK. When p>l the

system will be referred to as wideband MFSK. It has been shown that using values of p> 1 increases

the multiple-access efficiency by whitening the co-channel interference PSD at the receiver [22].

Since p is an integer, the symbols are still orthogonal and noncoherent detection is still possible.

Figure 4 depicts this system in a muLt=ple-access context, with N one way transmissions. Each of the

N transmitters spreads its signal with Lts own unique PN code, labelled PN_ through PAIN• Transmitter

III. MFSK Spread Spectrum Multiple Access 14
_"' > >1 Mps_,

[, _00.,

PH.{t}

/

S_

PN.(t )

J

.I

7

Figure 3. SSMA System Model

III. MFSK Spread Spectrum Multiple Access 15
j intends to communicate with receiver j. At the input to receiver j are N user signals plus thermal

noise. When the receiver correlates the j_' transmitted signal with a properly synchronized version

of the j_' PN code, the wideband MFSK signal si is recovered. Added to the recovered information

signal is a co-channel interference signal which consists of the correlation of the j_' PN code with N-1

signals spread by N-1 different, unsynchronized PN codes.

Up to this point, nothing has been said about the type of PN spreading code to be used. Though it

will not be discussed here, the co-channel interference caused by an asynchronous interfering user

is dependent upon the partial cross-correlations between the two PN codes [23]. Therefore it is

desirable to choose a family of codes with a low cross-correlation between any two members, such

as the well known Gold or Kasami codes. Since the number of interfering users will be large, and they

will all be transmitting asynchronously, a central-limit theorem argument can be used to model the

co-channel user interference as a Gaussian process [24]. Doing so gives an ensemble average

measure of the system performance. In practice some channels will exceed the average

performance, and others will fare worse. Making this Gaussian assumption reduces the performance

analysis of this system to that of wideband noncoherent MFSK in colored Gaussian noise.

Performance Analysis

Assume that Gaussian noise exists wIth a PSD given by/( f). The form of the MFSK signal is

olo

2_-_'_roct_t_jTs) cos( 2_rfl t ) i _ [1,M] (3.3)

III. MFSK Spread Spectrum Multiple Aces== 1S
xttt_ >

iI.,sK "°° !I

• PN I

• j

_oo o_ooj

PN

PN.

• J

i.oo J (_

OM

N

Figure 4. SSMA System Model. N Umre

III. MFSK Spread Spectrum Multiple Acctm 1 ?
Input

co82_rft

sin2_rft

Max Imum

Detector

Output

sin 27rft

Figure S. MFSK Receiver Model

III. MFSK Spread Spectrum Multiple Acce== iS
Figure 5 shows the model of the MFSK receiver. Letting sp denote that an MFSK symbol at frequency

fp was transmitted, the probability that the receiver incorrectly chooses symbol q given that symbol p

was sent is given by [Appendix A]

/Vq exp _ Es+ Pr (_ > _lsp) = /_ + Nq Nq

(3.4)

whereE, is the symbol energy (=kC_) and N_=/(f,), Nq=/(fq).

not made in this decision is

The probability that an error is

Pr(_<$ls o) = 1-Pr(_>_olsp) (3.5)

The probability that no errors are made over all of the incorrect symbols is

H[I - Pr (_ > _ISp)]

0,,-1

q,_D

(3.8)

Therefore the probability of a symbol error grven symbol p is transmitted is

,- FI[,-

_,,=I

(3.7)

Since the interference is not white, the probability of a symbol error is

average symbol error rate is found by averaging over all of the symbols

symbol dependent. The

p,,=___S"

M--.,

D.u_ II -

q,lJO

H [I -- Pr (rq> rol sp)]}

(3,8)

The corresponding bit error probab,hty can be approximated by using the relation

III. MFSK Spread Spectrum Multiple &cx:aN

19
Poe = fv_"_l Pse (3.9)

All that remains to be done is to find the form of the PSD of the interference,/(f). An individual MFSK

symbol from an interfering user is spread twice by two different asynchronous PN codes (Figure 6).

The PSD of the resulting signal is centered at the symbol frequency fp and has a shape determined

by the convolution of the PSDs of the two PN codes. Using an envelope approximation to the PSD of

a PN sequence, the resulting interference PSD can be approximated by [Appendix B]

2CRc (3.10) /(f) = 2

4R¢ + 2(f_ f/_)2

Since there are N-1 transmitters and each is transmitting any of the M symbols with equal probability,

on average there are (N- 1)/M users transmitting each symbol. Assuming all of the users' signals

are uncorrelated and R,<< R=, the powers add, and including thermal noise (No) the expression

becomes

I(f)= No +

M

2Re

(NM1)C _ 4R¢2 2(f_fj)2 j=,l 4-

(3.11)

With a frequency spacing between MFSK symbols of zlf, the interference at the i _' symbol frequency

is

N,= /(f_)= N_+

M

(N - 1)C S

2R c

4Re 2 + ,r2,M 2( i -j )2

(3.12)

III. MFSK Spread Spectrum Multiple Acceu 20
fp

I::Nx PNy

Figure 6. Interference PSD Model

III. MFSK Spread Spectrum Multiple Acceu 21
Multiple Access Systems

Using the described system to spread the energy of a low data rate MFSK signal over the entire

bandwidth of a satellite transponder is impractical for several reasons. First of all, the high chip-rate

needed to spread the bandwidth to 36 MHz or more requires expensive hardware. A high chip-rate

also greatly complicates the code synchronization problem over low chip-rate systems. Two alternate

methods of spreading the bandwidth in the channel will be presented. In both cases part of the

channel bandwidth must be allocated to a control channel. A simple control station can use this

channel to limit the number of users to a preset maximum and to assign PN codes to users when they

initiate communication.

System 1: MFSK/DSIFDM

One possible multiple access technique is to divide the available channel bandwidth into several

"FDM" slots, each containing a separate MFSK/DS SSMA system. The approximate bandwidth of one

slot is (Fig 7)

(M - I) _1/ + 2R¢ (3.13)

If the number of FDM "slots" that can fit in the channel bandwidth is denoted by Z, the number of

allowable users is simply Z times the number of users able to use one slot as given by equation (3.9).

III. MFSK Spread Spectrum Multiple Accltm 2 2
System 2: MFSK/DSIFH

Figure 8 shows a block diagram of hybrid $SMA transmitter and receiver which uses a frequency

hopped carrier tospread an MFSK/D$ signal over the transponder bandwidth.

Only slow hopping systems, with one frequency hop per symbol time will be considered. In a

conventional slow-hop FHMA FSK system, if two (or more) users hop into the same FH slot

simultaneously an error will occur (except for the case where all users are transmitting the same

symbol). Fast-hopping (multiple hops per symbol) eliminates the certain occurence of a symbol error

from a single collision. In a sense the hybrid $$MA system behaves like a fast-hop system, in that a

single FH collision does not necessarily cause a symbol error. The PN codes provide the

differentiation between intended and interfering .users.

The performance of the hybrid system can be calculated by applying the total probability theorem:

N

Pbe ==S Pr (bit errorl n users colliding) Pr (n users colliding)

nil

(3.14)

where N == the total number of users

If the user hops randomly among Z frequency slots, the probability that the user is in a given slot is

1/Z, and the probability that n users out of N are using that slot is given by the Bernoulli trials formula

Pr(nuserscolliding)=tNt I--_]n II--_.]N--n (3.15)

The probability of a bit error given n ,nterfering users is calculated using formula (2.14) with

Pr (bit errorln users colliding) calculated using equation (3.9) with N replaced by n in the interference

formula (3.12).

III. MFSK Spread Spectrum Multiple Acce== 23
... - ... • "...

° _" V

ae"

L_ ( M- 1)=31=* =R e :

Flgure 7. MFSK Slot Bandwidth

III. MFSK Spread Spectrum Multiple Accu= 24
i

I'°°

• PNI

OEMOO J

PNj

Figure 8. Hybrid MFSKJDS/FI'I SSMA System

III. MFSK Spread Spectrum Multiple Access 25
Numerical Examples

The following tables list the parameters of sample C and Ku band satellite SSMA systems. The

tabulated parameters are:

M

R,

R=

Z

the number of MFSK symbol frequencies ( = 2k)

the symbol rate (=R_/k)

the PN code rate

the frequency spacing between symbol frequencies

( = p R,)

the number of frequency hop slots in the FH system or

the number of FDM slots in the FDM system

III. MFSK Spresd Spectrum Multiple Acce-= 28
Example 1: C Band System

Transponder Bandwidth = 36 MHz

Data Rate (Rb) = 1200 bps

p =46

BER = 10-I

R= = 31 R,

M R, R= _ Z

2 1200 37,200 55,200 277

4 600 18,800 27,600 300

8 400 12,400 18,400 234

18 300 9,300 13,800 159

Example 2: Ku Band System

Transponder Bandwidth = 54 MHz

Data Rate (Rb) = 58 kbps

p=22

BER = 10-=

R= = 15 R,

R,, R=, &t" are in kbps and kHz

M R, R= _ Z

2 58 840 1.232 18

4 28 420 818 20

8 18.68 280 410 68 15

18 14 210 308 10

Figures 9-12 show the performance both multiple access systems for the C and Ku band

examples. The choice of parameters _s not optimized. Considerable flexibility exists in the

choice of parameters and considerations such as hardware complexity and cost may dictate

certain choices. The above examl:les serve only to illustrate the general behavior of the

systems" performance, t_f is an ,nteger multiple of R, and is set to approximately 1.5 times R c

since this choice empirically g_ves good I_erformance

III. MFSK Spread Spectrum Multiple A_:c_u 27
3S00 -

3250

3000

2750

25(_0'

2 2S0_

M :'16

M:8

20 0(3,

USERS.

1710-

1500.

1250.

000

750.

M=;

10 12

E_//No [riB]

14 16 18

Figure 9. Performance of MFSK/DS/FDM Systems (C band)

Ill. MFSK Spread Spectrum Multiple Accesl 28
3000

2750

250O

2250

20O

1750'

1S00.

USERS

1250,

1000

75(

500,

250'

M:16

M=5

I I I I I I I I I I I I I l I ! ! l I ! I I I I l I I !,,,,,,,, !

10 12 14 16 18

E_/N o_,[ dB ]

Figure 10. Performance of MFSK/DS/lrH Syma (C band)

III. MFSK Spread Spectrum Multiple Acailm 29
120"

110.

M=lS

100'

90' M: 8

80"

70

00

USERS

50, / / M: 2

40-

20

10 14

Flgurt 11. Performmnct of MFSKJDS/FDM Systems (Ku bend)

III. MFSK Spread Spectrum Multiple Acceu

30
M =16

USERS

4o

M =e

3o M=4

10

10 12

Flgum 12. Performance of MFSKJOS/FH Systems (Ku bend)

III. MFSK Spread Spectrum Multiple Access 31
Sample Link Analyses

Shown in Table 3.1 and Table 3.2 are link analyses for the C band and Ku band systems

described previously with earth station parameters chosen to be those of the common small

earth terminals. The network architecture is the "full mesh" type where direct communication

between two small earth terminals via the satellite, as in Figure 13, is assumed.

III. MFSK Spread Spectrum Multiple Accem 32
control station

Figure 13. Full Mesh Network Architecture

III. MFSK Spread Spectrum Multiple Acceso 33
Table 3.1. C-band System 968 users at 1.2 kbps

Satellite Parameters

EIRP = 36 dBW

G/T = -3 dB/K

Bandwidth = 36 MHz

Earth Station Parameters

Antenna Diameter = 1.2 m

Aperture Efficiency = 65

Transmit Gain (6 GHz) = 35.6 dB

Receive Gain (4 GHz) = 32.2 dB

Rb = 1200 bps

T, = 170 K

Down/ink

Propagation and pointing loss = 196.3 dB

Earth Station G/T = 9.1 dB/K

( C/No )_ = 45.0 dB-Hz

Upink

Propagation and pointing loss = 200.2 dB

Earth station power = 3.65 dBW

( C/No ), = 64.7 dB-Hz

Overall Link

Carrier to Interference

and Intermodulation Ratio = 18 dB

Overall Eb/No = 12.2 dB

BER = lOS

Throughput = 3.2

III. MFSK Spread Spectrum Multiple Accela 34
Table 3.2. Ku-band System - 57 users at 56 kbps

Satellite Parameters

E1RP = 38.5 dBW

G/T = 2 dB/K

Bandwidth = 54 MHz

Earth Station Parameters

Antenna Diameter = 1.8 m

Aperture Efficiency = 65

Transmit Gain (14 GHz) = 46.6 dB

Receive Gain (11 GHz) = 44.5 dB

R_ = 56 kbps

T, = 370 K

Downlink

Propagation and pointing loss = 206.8 dB

Earth Station G/T - 18.78 dB/K

( C/N, )_ = 61.5 dB-Hz

Uplink

Propagation and pointing loss

Earth station power

( C/N, ),,

= 208.5 dB

= -3.0 dBW

= 65.7 dB-Hz

Overall Link

Carrier to Interference

and Intermodulation Ratio

Overall Ea/N e

BER

Throughput

= 20 dB

= 11.9 dB

= 10-_

= 5.9

IlL MFSK Spread Spectrum Multiple Acces= 3S
Theavailabledownlinkpowerlimits the number of users to 968 for both the FDM and FH systems.

The Ku band system, with its higher antenna gain, is power limited for the FDM system to 57 users,

but is bandwidth limited for the FH system to approximately 43 users (Figure 11).

To determine the effect of such networks on adjacent satellites, the flux density they present to

adjacent satellites will be compared to that produced by a typical large earth station with a 9 meter

dish antenna (which meets the new FCC antenna gain specifications) transmitting 27 dBW. Such an

earth station presents a flux density of approximately -112 dBW/m z to a satellite at 2° off axis.

Assuming a uniform aperture illumination, at 6 GHz a 1.2 meter antenna's gain is down 6 dB relative

to its maximum gain at 2 ° off axis. The flux density of the C band system (with 968 users transmitting

simultaneously) at an adjacent satellite is approximately -100 dBW/m z. Reducing the flux density to

-112 dBW/m z limits the network to only 81 users. Alternatively, the diameter of the earth station

antenna can be doubled to 2.4 meters. This bnng the antenna gain down 17 dB at 2° off.axis, reducing

the interference flux density to-111 dBW/rn _ with no reduction in the number of users. At Ku band the

adjacent satellite interference problem is much less severe. At 14 GHz, a 1.8 meter antenna's gain

is down 24 dB at 2° off axis.

The wide beamwidth of these small antenna= guarantees that signals from adjacent satellites will

interfere with the network operation. In the link analysis, conservative carrier to interference ra¢io

values are included for this reason. Sprea¢l spectrum systems, unlike other multiple access systems,

have the ability to reject interfering s,gnals due to their processing gain. This capability, and the

relatively fiat PSD presented to adjacent satelhtes as interference, are the major advantages of spread

spectrum multiple access systems.

38

III. MFSK Spreld Spectrum Multiple J_.ce_
Conclusions

Both the MFSK]DS/FDM and MFSK/DS/FH SSMA systems provide reasonable multiple access

performance. Considering bandwidth constraints only, the FDM SSMA system performs better than

the FH SSMA system, providing bandwidth efficiencies of approximately 11 ( 0.11 bps/Hz ) for the C

band examples (1200 bps), and for the Ku band examples (56 kbps).

Downlink power limitations prevent the full capability of these systems from being realized in the

satellite 5SMA context. Considering both the power and bandwidth constraints present in the

example systems presented, the C band system throughput was reduced to 3.2 , and the Ku band

system to 5.9. Since these systems are essentially power limited, coding could be used to trade

power for bandwidth to increase system performance if the additional cost is tolerable. The link

analyses show that a small earth terminal satellite network using wideband MFSK SSMA is feasible.

Other types of multiple access systems are power limited but do not have the interference rejection

capability and fiat PBD of a SSMA system and may therefore become interference limited.

Because of its superior bandwidth efficiency and much simpler and faster synchronization, the FDM

$SMA system is best suited for a practical SSMA small earth terminal network.

Ill. MFSK Spread Spectrum Multiple Aoc=m 37
IV. Spread Spectrum Overlay Service

Overview

Overlay service is the practice of adding or over/aying a usually low rate data signal to an existing

channel which is generally thought to be filled to capacity, This practice, a form of "frequency reuse',

can further increase the capacity of satellite transponders and provide new services and functions to

existing satellite users.

The primary consideration in the design overlay systems is that they do not interfere significantly with

the operation of the signal which is already using the transponder. This will be called the "existing"

signal. The existing signal can be any one of many types of satellite signals currently in use such as

analog television, FDM/FM telephone traffic, or digital data. These signals already experience

interference from adjacent satellites and terrestrial sources. The addition of the overlay signal to the

transponder introduces another source of _nterference with which the existing signal must contend.

The performance degradation experienced by the existing signal can be kept to an arbitrarily low level

by simply lowering the transmitted power of the overlay signal sufficiently. This obviously places

IV. Spread Spectrum Overlay Service 3S
severelimitationsonthe performance of the overlay link, lowering its carrier to noise density ratio

C/No. Theoretically, any Eo/No can be achieved by the overlay link, for a fixed transmitl, ed power C, if

the data rate of the overlay signal is sufficiently low, since Eo = C/Ra. The relationship between the

overlay signal power and data rate illustrates the general characteristics of overlay systems. The

overlay signal will be a low data rate signal transmitted at a very low power relative to that of the

existing signal. Just as the overlay signal interferes with the existing signal, the existing signal

interferes with the overlay signal. The only difference is that the overlay signal must reject a relatively

high level interferer. Spread spectrum signals, which have the ability to reject large interferers, are

a natural choice for the overlay signal.

The performance of the overlay system is also highly dependent on the type of existing signal being

used. This chapter compares the performance of several spread spectrum signals used as overlay

signals for three types of existing signals: 1) analog television, 2) FDM/FM/FDMA telephone, and 3)

digital SCPC signals. Link analyses are given to assess the feasibility of the overlay systems. The

analysis includes calculations of the improvement from forward error correction.

Comparison of Spread Spectrum Overlay Systems

Frequency Hopped Spread Spectrum

In general, spread spectrum systems fall _nto two categories: 1) direct sequence systems, where the

signal is spread by being mixed directly w_th a pseudo-noise sequence, and 2) frequency hopped

systems, where the signal's career frequency is changed in a pseudo-random manner. Combinations

of these two systems, such as the hyOrld sDread spectrum multiple access system presented in the

previous chapter, also exist. Direct te<luence systems are known as "noise averaging" systems,

IV. Spread Spectrum Overlay Serv_== 3 g
becausetheir interferencerejection,or processinggain,is achievedby spreadingthe interference

signal powerover a larger bandwidth,reducingits power spectral density. Frequency hopped

systems are known as "avoidance _ systems, because they achieve interference rejection by hopping

their carrier into areas of the spectrum where there is no interference, thereby "avoiding _ the

interference. Since the existing signal occupies virtually the entire transponder bandwidth, it appears

as wideband interference to the overlay signal. In essentially no portion of the transponder bandwidth

will a frequency hopped signal avoid the existing signal. Therefore, pure frequency hopped overlay

signals provide no rejection of the existing signal, and are not suitable for overlay service.

MFSK/DS Spread Spectrum Oveday

The MFSKJDS spread spectrum system proposed in Chapter 3 for multiple access has a direct

sequence component, and therefore achieves interference rejection through noise averaging. Since

the frequency hopping component of the hybrid system provides no interference rejection, the

discussion will be confined to systems in which an MFSK signal is simply spread by multiplication

with a pseudonoise sequence.

If a large number of MFSK tones are used, i.e. M is large, the energy of the signal will be spread fairly

evenly over the transponder bandwidth. This relatively fiat power spectral density is a desirable

feature in overlay signals, since it is the peaks in the signal spectrum which would cause the most

interference to the existing signal. Nevertheless, there are several disadvantages to an MFSK/DS

system. First of all, noncoherent detection of the MFSK is required. The synchronization and general

hardware complexity of a coherent MFSK/DS system would make this system expensive and

complicated. Noncoherent detection schemes suffer a 3 dB penalty in signal to noise ratio relative to

coherent detection schemes. Because the interference level is high, this penalty will have a strong

effect on the overlay system BER. Also, Lf many MFSK tones are used to flatten the overlay signal

PSD, then the spreading sequence b_t rate must be reduced to keep the overlay signal power within

IV. Spread Spectrum Overlay Service 40
thetransponderbandwidth.Reducingthechipratereducesthebandwidththeinterferenceis spread

over.Thislowersthe abilityof thesystemto rejectinterference.

Direct Sequence Overlay

From the preliminary analysis of the MFSKJDS and frequency hopped overlay signals, it is evident that

an ideal overlay signal would have a flat PSD, use coherent detection, and have as large a chip rate

as possible. A pure direct sequence spread spectrum system comes as close to this ideal as any

spread spectrum system can. Figure 14 illustrates this system.

Such a system achieves a high processing gain, using a chip rate up tohalf the transponder

bandwidth, and can easily be detected coherently, providing a 3 dB advantage over noncoherent

systems. While the PSD of the signal is not.as fiat as some other spread spectrum systems, the

advantages just mentioned more than make up for this disadvantage. This system seems to have the

most promise for overlay service, and will be analyzed exclusively,

Interference Modefing

Appropriate statistical models of the interference that the overlay signal presents to the existing

signal, and vice-versa, are an important part of the overlay system performance analysis. It will be

shown that with a reasonable degree of certainty these interference sources are adequately

represented as Gaussian noise. The following sections describe two results which are useful in

justifying this treatment.

IV. Spread Spectrum Overlay Service

41
OA TA

IN

P N fup

BI:F

f

fdown P N

OA TA

OUT

Figure 14. DS Overlay System

IV. Spread Spectrum Overlay Servia 42
Modeling of Wideband Signals Passed Through Narrowband Filters

The statistics of an arbitrary signal passed through a filter with a bandwidth much narrower than the

signal bandwidth approach a Gaussian distribution. This is a result of the central limit theorem and

the related Berry-Esseen theorem [25,26]. In the overlay systems to be analyzed the interference

bandwidth is often much greater than the a receiver's bandwidth. This result can then be used to

model the interference in the receiver as a Gaussian process.

Modeling of Interference PSK Signals by Multiple PSK Interferer=

In his classic paper V.K. Prabhu considers the error rate performance of a phase shift keyed signal

in the presence of Gaussian noise and co-char_nel interference consisting of K identical phase shift

keyed signals [27]. He shows that the error rate increases monotonically as K increases (with the

total interference power constant and equally distributed among the K interferers). As K approaches

infinity, the error rate reaches its maximum and is identical to the error rate calculated assuming the

total interference was white Gaussian noise. The significance of this is that for the practical case,

where K is finite, assuming that the interference is Gaussian noise is conservative: this assumption

overestimates the bit error rate. This result is directly applicable to one of the cases to be analyzed,

where the existing and oveday signals are phase shift keyed signals. One may argue that the worst

kind of Interference a signal may experience is from a signal of the same type, since the receiver

detects this signal just as well as the mtended signal. If using a white noise model overestimates the

error rate in this case, as Prabhu has shown, it should also overestimate the error rate for a

non-identical signal.

IV. Sprmld Spectrum Overlay Service

43
Overlay System Performance Analysis

Overview

The analysis of the performance of the overlay signals will be done in three steps. First the maximum

overlay signal power that can be transmitted from a small earth terminal (transmitting a BPSK overlay

signal) without causing significant interference to the existing signal will be determined. Next the bit

error rate performance of the overlay link must be calculated. This is done by determining the PSD

of the interference that the overlay signal receiver sees. Since the existing signal is spread at the

overlay receiver, the PSD is found by numerically convolving the PSD of the existing signal with that

of the spreading signal in the receiver. This PSD is used to calculate the overlay link carrier to

interference density ratio. Finally a complete link analysis for the overlay signal is performed. It uses

the transmitted power calculated in the firststep, and the carrier to interference density ratio

calculated in the second step. For a specified BER, a maximum data rate for the overlay signal can

be calculated. This will be the measure of' performance of the overlay system. The improvement

provided by various forward error correction codes will also be calculated.

Description of Existing Signals

As mentioned previously, the overlay system performance depends on the type of traffic existing in

the satellite transponder, the existing signal. Since all type of existing signals can not be considered,

the analysis will be limited to the three signals described below.

1) An FDM/FM/FDMA telephone signal consisting of eight 4 MHz channels in

an FDMA mode,

2) An SCPC system consisting of two hundred 64 kbps channels using QPSK

IV. Spread Spectrum Overlay Service

44
modulation,

3) A single FM analog television signal occupying virtually an entire 36

MHz transponder.

All signals are assumed to be transmitted from a "typical" large earth station with a 9 meter diameter

dish antenna, and approximately 200 watts of total power.

Overlay on FDMIFM/FDMA Telephone Traffic

Figure 15 shows the first existing signal to be considered. It consists of 8 FDMA channels with a

bandwidth of 4 MHz each. An earth station with 9 meter diameter antenna transmits each FDMA

channel with a power of 25 Watts.

Figure 16 shows the PSD of the overlay signal. It is a simple BPSK signal with a chip rate of 18 MHz.

The overlay earth station has a two meter diameter dish antenna. The transmit power is to be

determined. The analysis assumes a transponder bandwidth of 36 MHz and operation at C band.

The total bandwidth of the overlay signal is 36 MHz. Any one of the 8 FDM channels will receive only

a portion of the overlay signal power. The bandwidth of one of these channels, 4 MHz, is only one

ninth of the overlay signal bandwidth. The wideband overlay signal therefore passes through a

relatively narrowband filter before being detected by the FM receiver. As discussed previously, when

a wideband signal passes through such a filter the ouptut has Gaussian statistics. For this reason the

interference to any one of the eight channels of the existing signal caused by the overlay signal will

be modeled as Gaussian noise.

To calculate the maximum allowable transmitted power for the overlay signal, the power which causes

the signal to interference ratio in the worst case channel to be 25 dB must be determined. Since the

45

IV. Spread Spectrum Overlay Service
L.

L_ 38MHz J

Figure 15. FDMA Existing Signal

IV. Spread Spectrum Overlay Service 48
!..,, .36 MHz ' ,,'-|

r"

Figure 16. OS Overlay Signal

IV. Spread Spectrum Overlay Serv_:a 4 7
PSD of the overlay signal is at its highest in the center of the transponder, channels 4 and 5 in Figure

15 experience the most interference. The fraction of the overlay signal power contained in a 4 MHz

band around its center is approximately 22%. For the signal to interference ratio at the transponder

to be 25 dB in channel 4 or 5, the overlay signal power (again at the satellite) needs to be

-25 dB - 10 Iog(,22) =-19 dB

The overlay signal power at the transponder must be 19 dB below the power of a single FDMA channel

at the satellite. The FDMA signal has a transmitted power of 14 dBW ( =25 watts) and its earth station

antenna has a gain of 48.2 dB, giving it a total EIRP of 62.2 dBW. The overlay signal EIRP must be 19

dB below this, and since its 2 meter antenna has a gain of 40.1 dB, it must transmit 3 dBW, or 2 watts.

Now that a tranmitted power constraint has been placed on the overlay signal so that the performance

of the FDMA telephone system is not degraded appreciably, the performance of the overlay link must

be analyzed. The interference of the existing signal to the overlay signal must first be considered,

then a conventional link analysis using the parameters of the overlay link and earth station can be

performed.

As can be seen from Figure 14, the signal received by the overlay receiver is despread by mixing it

with a replica of the spreading signal, an 18 MHz chip rate BPSK signal. To calculate the signal to

interference ratio for the overlay link (the interference being the existing signal), the effect of this

despreading must be calculated. This is done by convolving the PSD of the existing signal with that

of the spreading signal, with the PSD of each FDM signal approximated by a Gaussian curve with a

standard deviation of 546 kHz [28]. Figure 17 shows the result, with the power of a single FDMA signal

normalized to 1.

The matched filter in the overlay recewer has a bandwidth approximately equal to the bit rate of the

overlay signal. If the overlay signal data rate _s on the order of several kilobits per second, and since

the bandwidth of the interference is at 'east as large as the spreading signal bandwidth of 36 MHz, the

statistics at the output of the matched fiiter are very nearly Gaussian. Since the overlay signal power

IV. Spread Spectrum Overlay Service 4@
Ul

:),

0

E

t;

Q.

"U

m

i,

__2
is 19 dB below that of the FDM signal, or 12.6 mW normalized, and the noise sDectral density at the

center of the transponder bandwidth is 2 x 10-_ Watts/Hz, the carrier to interference density ratio at

the overlay receiver is

0126 W --. 48.0 dB-Hz

2 x 10.7 W/Hz

The following link analysis combines th=s carrler to _nterference density ratio with the carrier to noise

density ratios of the upllnk and downlink to determine the link EolN, (Table 4.1).

IV. Spread Spectrum Overlay Servt<e

50
Table 4.1. FOM Overlay System: C- Band

Saterite Parameters

EIRP =

G/T =

Bandwidth =

36 dBW

0 dB/K

36 MHz

Earth Station Parameters

Antenna Diameter

Aperture Efficiency

Transmit Gain (8 GHz)

G/T

= 2.0 m

= 65

= 40.1 dB

= 14 dB/K

Downlink

Propagation and pointing loss

( C/N, ),

== 198.3 dB

== 47.9 dB-Hz

Uplink

Propagation and pointing toss

Earth station power

( C/N, )_

- 200.2 dB

= 3.00 dBW

== 78.6 dB-Hz

Overall Link

Carrier to Interference

Density Ratio

Overall C/N,

= 480 dB-Hz

== 44.9 dB-Hz

IV. Spfll_l SpecS'urn Ovede¥ Service

Sl
With an overall link carrier to noise density ratio of 44.9 dB-Hz, and a Ea/No of 11 dB needed to achieve

a 10-I bit error rate, the maximum bit rate achievable is

44.9 dB-Hz - 11 dB = 33,9 dB-bps

or

= 2475 bps 0 max

While this data rate is quite low, it can be increased by using forward error correction. The chip rate

can be held constant at 18 MHz regardless of the data rate by changing the length of the code. If the

chip rate is held constant, all of the interference PSDs and link parameters mentioned previously are

unchanged.

A rate 1/2, constraint length 7 Viterb_ decoded convolutional code gives a 8 dB improvement in E,/N.

at a bit error rate of 104 [29]. Therefore the achievable data rate =s increased by 8 dB by using th=s

coding scheme. With this coding:

R o max = 9855 bpS

Overlay of Digital SCPC System

Figure 18 illustrates the digital SCPC system to be considered. Lt consists of 200 QPSK signals at 64

kbps each transmitted at I wart ,nrougn a 9 meter diameter earth station, frequency diwsaon

multiplexed in a 38 MHz transponder

Nyquist filter is approximately 45 ,Pz

"he RF bandwidth of each of these signals With a 50% rolloff

The overtay s_gnal is unchanged from the previous section.

s2

IV. Spread Spectrum O_er_rr _14r_¢=
.J

I., 3e MI'I Z _°

Figure 18. SCPC Existing signal

IV. Spread Spectrum Overlay Service $3
Using the same arguments as before, the statistics of the interference at the SCPC and overlay signal

receivers will be treated asGaussian. The 36 MHz bandwidth of the overlay signal is much wider than

that of the SCPC receiver, causing the interference at the SCPC receiver to have Gaussian statistics.

Furthermore, the multiplication of the 200 SCPC signals by the despreading signal in the overlay

receiver creates an interference signal with a bandwidth much larger than 38 MHz, which is much

larger than the bandwidth of the overlay receiver. Therefore the interference to the overlay system

is also Gaussian.

The analysis of the overlay system performance is identical to that for the FDM system. The fraction

of the overlay signal power contained in the bandwidth of any SCPC signal is less than 1%. One

percent will be used as a conservative figure. Therefore for the signal to interference ratio in the

bandwidth of one SCPC signal to be greater than 25 dB, the overlay signal power relattve to the power

of an SCPC signal (at the satellite) must be -25 - 10 log (.01) = -5 dB. This gives an overlay signal

transmit power of 2 Watts. Performing a numerical convolution to determine the interference P_3D at

the overlay receiver, with the narrowband SCPC signals approximated by impulse functions in the

frequency domain, gives an interference spectral density of 5.5 x 10-6 Watts/Hz (see Figure 19). This

gives a carrier to interference density ratio of 47.60 dB-Hz. The complete link analysis follows (Table

4.2).

IV. Spread Spectrum Overlxy Service

54
io-_6 o.

$,S

SO

4.S

4.0"

3.S'

PSD

3'0 1

2.1

2.0'

1.6

OOo ' 1_ 2o' ;o ,'o " s'o e'o

F CM-- ]

'l

70

Figure 19. Interference PSD caused by the SCPC signal

IV. Spread Spectrum Overlay Service $Ii
Table 4.2. SCPC Overlay System - C Band

Sate/rite Parameters

EIRP =

G/T =

Bandwidth =

36 dBW

0 dB/K

36 MHz

Earth Station Parameters

Antenna Diameter

Aperture Efficiency

Transmit Gain (6 GHz)

G/T

= 2.0 m

= 65

= 40.1 dB

= 14 dB/K

Down/ink

Propagation and pointing loss

( C/N, ),

= 196.3 dB

= 54.3 dB-Hz

Up/ink

Propagation and pointing loss

Earth station power

( C/N, )=

= 200.2 dB

= 3.1 dBW

= 76.6 dB-Hz

Overall Link

Carder to Interference

Density Ratio

Overall C/No

= 47.6 dB-Hz

= 48.75 dB-Hz

IV. Spread Spectrum Overlay Servtce

Sl
ORIGINAL PAGE IS

OF POOR QUALITY

Again since an 11 dB E_/No is needed to achieve a 10-e BER, the maximum data rate of the overlay link

is 3758 bps. Using a rate one-half convolutional code with a coding gain of 6 dB increases the data

rate to approximately 15 kbps.

Analog Television Overlay

The overlay of data signals on analog televis=on traffic presents some unique problems. First of all

the spectrum of the television signal is not known with any degree of certainty. Its shape varies

significantly with the type of image being sl_own. Secondly, in contrast to the FOM and SCI:_ cases,

the television signal is .a wideband signal. Essentially the entire overlay signal will be contained

within the bandwidth of the television receiver, not only a small portion as is the case with the

relatively narrowband FDM and SCPC reclevers. Finally, the degree to which an interferer degrades

the performance of a television signal is higmy subjective, and cannot be quantified. Television is

primarily a video signal. Even though an _nterferer may have a very low power relative to a television

signal, if it has certain characteristics. =t may produce objectionable images in the received television

video. Furthermore, high level interferers wtth the right parameters may produce little noticeable

interference. For these reasons, telews=on interference research is best performed by experiment

with human judges. Computer simulat=on may be useful, but will still fail to include the subjective

measures of video quality that are an _ml)ortant part of the system design. This section presents

estimates of the acceptable interference levels to provide a crude estimate of the feasibility of signal

overlay on analog television signals

In satellite television transmiss=on tr_e oaseOand television signal, with a video bandwidth of

approximately 4 MHz, is FM modulated _o treat ,ts RF bandwidth tills an entire 38 MHz transponder.

When the overlay signal, which is a 8P_, _,gnal w=th a data rate of 18 MHz. passes through an ideal

FM demodulator, the output would a_oear at ,n Figure 20, an impulse at each bit transition. The

IV. SpreId Spectrum Overlay Servk:_ S7
OR_G|_AL PAGE _$

OF POOR QUALFfY

impulses would be of random polarity and occur at a maximum rate of 18 MHz. The video recetver

is essentially a 4 MHz wide filter. The fact that this wideband (> 18MHz) noiselike signal passes

through a filter that has a relatively narrow bandwidth supports the assumption that the amplitude of

this interference signal at the output of the video receiver has Gaussian statistics. For this reason _t

is assumed that the BPSK overlay signal should not produce especially objectional interference to the

video signal. Since its distribution should be a nearly Gaussian function, it should not be more

objectional than white noise of equal power.

Since the entire overlay signal power is within the television receiver bandwidth, to keep the signal

to interference ratio at 25 dB the EIRP of the overlay transmitter should be 25 dB below that of the

television transmitter. If we assume a television earth station with a 9 meter diameter antenna

transmitting 200 Watts, the overlay transmit power from a 2 meter antenna should be approximately

4 Watts.

If we assume that the PSD of the telews_on signal can be approximated by the following Gaussian

curve [30]

I exp[ (f- 2o=fc)2 ]

v 2_a

where _ = 7.6 MHz, then the numerical evaluation of the interference PSO at the overlay receiver is

shown in Figure 21.

The carrier to interference density ratto ,s then 49.32 dB-Hz. The analysis of the overlay link is given

below (Table 4.3).

Iv, Spread Spectrum Overlay Serv,ce Sll
BPSK

INPUT A_

Sl

FM DE TE¢. TO R l

OUTPUT

--t

Flgure 20. BPSK $1gmll Passed "l"_ro_IIh _m _ Demodulator

IV. $prlsd Spectrum Overlay Servtco 59
o

IO
Table 4.3. Analog Television Overlay - C Band

Satellite Parameters

ElRP = 36 dBW

GtT = 0 dB/K

Bandwidth = 36 MHz

Earth Station Parameters

Antenna Diameter = 2.0 m

Aperture Emciency = 55

Transmit Gain (6 GHz) = 40.1 clB

G/T = 14 dB/K

Downlink

Propagation and pointing loss = 198.3 dB

( C/N, ), -- 57.8 dB-Hz

Uplink

Propagation and pointing loss :, 200.2 dB

Earth station power = 8.1 dBW

(C/N,). = 74.sdS-Hz

Overall Link

Canter to Interference

Density Ratio = 493 dB-HZ

Overall C/No = 487 clB-Hz

IV. Spread Spectrum Overlay Se_=ct

ill
ORIGINAL PAGE IS

OF POOR QUALITY

The overall link C/N. of 48.69 dB-Hz implies a maximum overlay data rate of 5874 bps at a 10-I 8ER.

The 6 dB coding gain from a rate 1/2 convolutional code increases th_s data rate to approximately 23.5

kbps.

Extension to MultJple Overlay Signals

In the case where N simultaneous overtay s=gnals are desired, the maximum data rate of each must

be reduced by a factor of N. For example _f the maximum allowable data rate for a single overlay

signal is 10000 bps, then then two overlay signals of 5000 bps, or 4 signals of 2500 bps, etc., can be

accomodated.

The reasoning for this is as follows, if the number of desired signals is N, then to keep the total

interference power constant when N signals are used, the power of each must be reduced by a factor

of N. This would lower the Eo/No of each s_gnal by a factor of N. To __ornpensate for this decrease to

keep the BER at the same level, the data rate must also be reduced by a factor of N to keep E=IN,

constant.

Conclusions

This chapter demonstrates that overtay of spread spectrum signals on existing satellite traffic is

feasible. The most promising spread sOec_rum modulation for this purpose appears to be ordinary

direct sequence spread spectrum _v_,fe :_,s _pe of service is poss=bte, the modest data rates it is

able to support suggest that this serv,ce ,'nay be _mpractical. Data rates of tess than 20 kbps are

generally insufficient to accomo<late a s_ngle vo=ce channel, unless sophisticated bandwidth

IV. Spread Spectrum Overlay Serwce e2
compression schemes are used. However, these data rates can support the operation of several low

data rate computer terminals.

IV. Spread Spectrum Overlay Serv_.o 63
V. On Board Processing for Multiple Access Systems

Introduction

All of the systems considered so far and probably all commercial satellites in existence use what is

known as a "classical transponder" The classical transponder amplifies the received signal and

translates the uplink frequency to the al_l_roprlate downlink frequency (Figure 22). While this

arrangement has proven itself to be extremely reliable, it has one disadvantage: the satellite

retransmits the uplink noise along with the ulolink signal increasing the noise received at the earth

station. Satellites with "processing trans_onclers" can decouple the effects of the uplink and downlink

noise. All of these not only amplify the received signal, but detect it. regenerate the best estimate of

the transmitted bit stream, and remodulate the data at the downlink frequency, as shown in Figure 23.

In systems where the uplink and downlink E_/No are nearly equal, this type of system can provide

about 3 dB gain. For instance, if the uolinK and downlink E_/N o for a conventional system are each

10.6 dB, the overall link Ea/N o determined from the formula

1 (5.1)

V. On Board Proceuing for Multiple Access Systems 84
Figure22. CllutcidTransponder

V. OnBoardProcessingforMultipleAccessSystems (iS
Figure 23. Procemdng Transponder

V. On Board Processing for MultIDte AccQN Symems (ill
is 7.6 dB. For small error rates, the following formula approximates the link bit error rate when a

processing transponder is used [31]:

Pe = (Pe) u_ +' (Pe) c'ow. (5,2)

At an E=/N, of 10.6 dB, P, for the uplinl< and downlink are each 10-= , making the link 8ER

approximately 2 x 10 -= . The link E=/No necessary to achieve this BER without a processing

transponder is approximately 10.3 dB. The i0tt error rate achieved by the processing transponder with

an overall link E=/N= of 7.8 dB is identical to that achieved by a conventional transponder with a link

E=/N, of 10.3 dB, a gain of 2.7 dB.

The previous example is not typical. In many applications the uplink and downlink E_N e are very

different, meaning that there is little contribution to the overall noise by either the uplink or downlink.

In these cases the benefit of a processing transponder is slight.

Processing transponders need not transmit the downlink signal with the same type of modulation as

the uplink. In some cases, changing the modulation can prowde advantages such as lower

interference, reduced hardware comptexmty, and a lower bit error rate. depending on the application.

One commercial satellite system that may be used in the future has an FDMA uplink transformed by

a processing transponder to a TDM downt=nt(. This system retains the advantages of FDMA (no tirr{e

synchronization between users and therefore s=mple equipment) while eliminating the need for any

backoff of the satellite power amphfler since mtermodulation interference is not a problem in TDM

(32].

In the following sections, the techmques descn0ed here are applied to the multiple access problem

described in Chapter 3 to determine ,f an(| wr_at type of processing transponder would increase the

throughput of multiple access from small eanr_ termtnals.

V. On Boerd Proce_ing for Multiple A_:_e= S_,_ S7
Simple Processing Transponder

As mentioned above, simply decoupling the uplink and downlink has little effect if their signal to noise

ratios differ appreciably. Unfortunately this is the case for the spread spectrum multiple access

systems considered in Chapter 3. Looking at the link analysis in Chapter 3 for the (3 band multiple

access system with 968 users, the uplink C/N, is 64.7 dB-Hz while the downlink C/N o is 45.0 dB-HZ.

The downlink noise determines the link performance. In effect, the difference in C/N o , since it virtually

negates the effect of the uplink on the link performance, has already decoupled the uplink and

downlink. The same is true for the Ku band system. Both systems are downlink limited. The effect

of a simple processing tranponder on the system performance would be minimal.

Dual Modulation Systems

While simply demodulating the received spread spectrum signals and retransmitting spread spectrum

signals would provide little advantage, there are benefits in using non spread spectrum modulations

for the downlink. Since the links are downlink limited, anything that would lower the downlink noise,

and therefore increase the downlink C;No would increase the system performance significantly. One

obvious source of noise that could be el=mfnated if a non spread spectrum modulation is used is the

inter-user interference. Much of the down[=nk noise is simply the interference between different users,

which are forced to share the same banawrdth at the same time.

Using a spread spectrum uplink is destra0te to reduce adjacent satellite interference. On the uplink.

the relatively fiat PSD of a spread s;ec',r_m ;;gnal would =nterfere little with adjacent satellites. As

V. On Board Processing for Multiple Access Sy_em= 61
explained in Chapter 1, this is desirable since the wide beamwidth of the small earth terminars

transmitting antenna would transmit significant power in the direction of adjacent satellites.

While interference to other users is not a problem for the downlink, since the antenna beamwidth is

fixed by the desired coverage area on the earth, the use of non spread spectrum modulations would

reduce the interference rejection capabili_ of the small earth terminal receiver. Since the wide

beamwidth of a small earth terminars receiving antenna will receive signals from adjacent satellites,

some form of rejection of these signals is necessary. Obviously the advantages of a transponder

which changes the modulation of the downlink signal can only be realized when the means exist to

reduce this interference to an acceptable level.

If the processing gain of a spread spectrum receiver is not available to reject interfering signals, the

only means to reject these signals is with the directional gain of the antenna. This immediately

excludes the use of non-spread spectrum downlink signals from C band full mesh VSAT networks.

The gain of a small dish antenna at several degrees off axis is only reduced by 4 or 5 dB. Signals from

adjacent satellites are therefore at a relatively high level, causing severe interference.

One exception to the use of such a system at C band is in multipoint to point networks, as shown in

Figure 24. Here many small earth terminals communicate with a single large earth terminal, which

may be connected to a central computer o4"telephone network. Because a narrow beamwidth (large

diameter) antenna can be used for _eptmn, signals from adjacent satellites are attenuated

sufficiently.

Small earth terminal networks using s_read spectrum uplinks and non spread spectrum downlinks

are feasible at Ku band and a0ove ,v_ere a small diameter antenna can provide significant

attenuation of signals from adjacent _atemtel One such system follows.

V. On 8o=rd Procemdng for Mu_pkl A4:z:_m $yltm_l 6S
cornl_l e r _

Figure 24. MuWpoint to Point Hetwort Arc_fteclura

V. On Board Processing for Multiple A¢ceee $vgmme 70
MFSKITDM Processing Transponder System

The Ku band spread spectrum multiple access system presented in Chapter 3 is limited by the

downlink carrier to noise ratio. To improve this ratio, which is influenced significantly by the

co-channel interference that the users present to one another, an appropriate non-spread spectrum

multiplexing scheme must be chosen. Any type of frequency division multiplexing would require that

the satetlite's transmitted power be backed-off to reduce intermodulation interference. Since the

satellite demodulates atl of the users" signals, the synchronization required in TDM systems can be

done at the satellite. Since no syncl_ronlzation Detween different transmitters is needed, a TOM

downlink signal should give the best system performance.

Figure 25 shows the operation of the on-OoarO processing system described, N parallel receivers

demodulate the N MFSK/DS spread spectrum s=gnais each with a data rata of R=. Each synchroniZeS

to one of the transmitted signals. A baseOand digital multiplexer accept= the data from each of the

receivers and forms a single data stream at a rate N x Rj,. This TDM data stream is then transmitted

at the downlink frequency using BPSK modulation. The earth stations then demodulate the BPSK

signal. At this point, all of the data sagnals from each transmitter are available at every receiver.

The analysis of the performance of th0s system _s clone separately for the uglink and downlink (Table

5.1). The link bit error rate is given t_y equation 5 2.

V. On Board Processing for Multtpte Accm Svstema 71
Table 5.1. Ku Band MFSK/DS - TDM System :110 users

Satellite Parameters

EIRP = 41.0 dBW

G/T = 2 dB/K

Bandwidth = 54 MHz

Earth Station Parameters

Antenna diameter = 2.0 m

Aperture Efficiency = 65

EIRP = 46.8 dB

G/T = 18.8 dB/K

Up/ink Parametera

Propagation and Pointing Loss

Uplink E=JNo

= 208.5 dB

= 21.2 dB

Oownlink Parameters

Propagationand pointing loss

TOM bit rate

Downlink E_/N,

206.8 dB

8.18 Mbps

13.7 dB

V. On Board @tor.assingfor Multiple A¢_ Synm4 ;'2
Thelink analysis confirms that this system can accomoclate 110 users at 56 kbps versus 57 users for

the same system which uses a classical transponder. The throughput has been increased by 93% from

.059 bps/Hz to .114 bpslHz. The system is now limited by the uplink to 110 users assuming that M= 16

and the MFSK/D$/FDM multiple access system (Figure 11) is used. The downlink E_/N, is 13.7 dB

which gives a downlink BER of less than 10-' The MFSK/DS/FDM multiple access system can handle

110 users for M=18 at an E_/No of 18 dB. Since the uptink E_JNo is 21.2 dB, its BER will also be much

less than 10-', assuring that the system BER is less than 10-= .

The throughput of the system is nearly doubled. The price for this increase in performance is an

extremely complex and expensive satellite, more complicated than probably any in current

commercial use.

V. On Board Pt,ocemdng for Multiple Acce,m Sysleml

73
Vl. VSAT Networks - An Overview

This section examines the technology and constraints of very small aperture terminal (VSAT)

networks, a special type of wide-area thin-route satellite network that represents a recent innovation

in the field of satellite communications. VSAT network architectures suitable for both data and voice

communications are studied in this chapter. Several issues concerning the frequency of operations

that is, C-band versus Ku-band are examined, and trade-offs between non-spread spectrum and

spread spectrum techniques, as well as modulation and multiple access schemes, are considered in

detail. Link design examples are given to illustrate the performance of various types of VSAT

networks,

Introduction

The vew small aperture terminal (VSAT) (or micro earth terminal) is a low-cost satellite earth

station with a small dish antenna and low power transmitter that is already playing a major rote in

data communication networking. VSAT networks provide viable alternatives to bypass terrestrial

telephone systems by bringing a numl:er of inherent advantages to the data communication problem,

VI. VSAT Networks - An Overview 7 4
suchas distanceinsensitivity,higher data rate (in the range of 1.2 kbps to 128 kbps) and good

reliability. In this section, we look at this important bypass technology from the system point of view.

Satellite communications offers unique advantages for networks which have a very large

number of remote terminals which occasionally communicate with a hub station. Additional remote

terminals can share the communications medium at little additional cost. However, to make such

networks viable, the VSAT used at each remote location must have a low capital and operational cost

- typically a few thousand dollars per unit compared to the millions of dollars required to purchase

and operate a large earth station. In VSAT networks, the capital cost of the earth station dominates

the economics of the system.

The next four sections discuss various VSAT architectures, the critical differences between 8/4

. GHz (C-band) and 14/12 - GHz (Ku-band) operation, the power-limited nature of VSAT networks and

link analysis for non-spread spectrum VSAT networks that are compared to spread spectrum VSAT

networks, and a look into the future of VSAT technology.

Architectures

VSAT networks are wide-area, thin-route satellite networks. They may have a variety of

architectures depending on the applications: single-hop, double-hop, a mixture of both single-hop and

double-hog (hybrid), and full mesh. The single-hop architecture (Fig.25) connects many remote VSATs

to a hub station at a central site or headquarters. The communication links are the remote-to-hub link

and the hub-to-remote link. The hub station may be connected to the terrestrial telephone or data

network via a circuit switch or a packet-switched network via a packet switch. Both data and voice

communications are possible via a single-hop link. Several data terminals or personal computers

may be connected to a remote VSAT which may also be able to handle one or two digital voice

channels at 32 kbps (using adaptive delta modulation or adaptive differential PCM). A VSAT network

like this one may have hundreds or even thousands of widely distributed data terminals, each

VI. VSAT Network= .. An Overview 7S
generating a low duty cycle (bursty) data traffic communicating with a host computer connected to the

hub station.

In the double-hop architecture, the remote VSAT communicates indirectly through the hub

station which is capable of switching calls originating from one remote VSAT to a second remote

VSAT. Because a link experiences a double-hop delay, this architecture is suitable for data traffic but

not for voice traffic (unless it is recorded voice traffic).

A mixture of both single-hop and double-hop architectures (Fig.26) provides a general and

attractive configuration for many applications. In this hybrid architecture, connectivity between the

hub station and the remote VSAT is possible (voice, data) as well as the connectivity between remote

VSAT (data and recorded voice messages). In the lull mesh architecture (Fig.27), any remote VSAT

may communicate directly with any other remote VSAT. A master control station must be used to

control the network activity and assign channels on demand.

The selection of a VSAT architecture depends on the type of networking. The single-hop

architecture in Fig.25 is the simplest and perhaps the most widely used network for this start-up

technology. The full mesh architecture in Fig.27 is perhaps the most complex but offers full

connectivity and may use the satellite capacity more efficiently in a demand assignment mode.

Ku-Band versus C.Band

VSAT networks are now available at both the 6/4-GHz and 14/12-GHz bands. The first

commercial VSAT system operating _nthe 6/4-GHZ band was introduced by Equatorial Communication

Company in 1981 [33,35]. This system uses spread spectrum technology to greatly reduce the power

flux density of the transmitted signal and thus avoid interference to adjacent 6/4-GHz satellites and

6/4..GHz terrestrial microwave systems The spread spectrum signal is spread by a pseudo-noise (PN)

code (or sequence) over a satethte bandwidth that is much wider than the data bandwidth. The

receiving equipment correlates the recewed signal with the same PN code to extract the transmitted

VI. VSAT Netwerkl - An Overview 7 s
I UOST

J Host

[coM,_I

Flgure TS. Single.Hop Architecture

V1. VAST Network--An Overview 77
Flgure 28. Hybrid Architecture(Mixture of Single.hop and double.Hop)

VI. VAST Network---An Overview 70
_,ConCro_ Station

Flgum 27. Full Mesh Archttecturt

VI. VAST Network_An Overview 79
ORIGINAL PAGE IS

OF POOR QUALITY

data. The Equatorial VSAT network uses the scngle-hop architecture. Even with spread spectrum

technology, the severe interference limitations at the 6/4-GHz band can cause any spread spectrum

VSAT network to use the satellite capacity inefficiently, and consequently the link data rate is low.

The average efficiency for a 8/4-GHz VSAT network is estimated to be less than 0.03 bps/Hz and the

average data rate is less than 9.6 kbps. The inefficiency of spread spectrum VSAT networks at the

6/4-GHz band has prompted unfavorable criticism [36]. Undoubtedly, spread spectrum systems use

transponder bandwidth inefficiently. But w,thout the use of spread spectrum signals, there could be

no 6/4-GHz VSATs with dish antennas as small as 08 meter in diameter, at a potential price of less

than $7,000 per terminal. It is likely that the market place will decide how the satellite bandwidth is

going to be used.

The situation is very much different for VSAT networks operating in the 14/12-GHz band which

is free of terrestrial microwave interference In the United States, the Federal Communication

Commission (FCC) has relaxed licensing requ=rements for relatively large VSAT networks (500

terminals) operating in the 14/12-GHz band in this band, VSAT networks can be designed to take

advantage of the higher power flux density allowed and the lack of terrestrial interference. They can

provide data rates of 58 kbps or more using dis11 antennas with diameter in the range of 1.2-1.8 m.

For the same antenna diameter, the antenna gains in the 14/12-GHz band are approximately 7.4

riB(transmit) and g.5 dB (receive) h_gner than those in the 6/4-GHz band. On the other hand, the

attenuation caused by heavy rain w,II be mucl_ greater at 14/12-GHz band than at 6/4-GHz band, and

consequently the VSAT networks oDeratsng at the higher frequency band suffer more and longer

outages. A study of the constraints for vSAT networks operating in the 14/12-GHz band has recently

been discussed in [37].

VI. VSAT Networks - An Overview 80
Power.Limited Networks

Because of the use of very small antennas and low power transmitters, a VSAT network is

inherently a power.limited network. Therefore, the satellite capacity usage is very inefficient whether

spread spectrum or non-spread spectrum technology is employed. This is obvious from the analysis

of three VSAT networks (Tables 6.1-6.4) us4ng PSKJQPSK as the modulation scheme and TOM and/or

SCPC as the multiple access technique. The analysis of these particular non spread spectrum VSAT

networks shows that the throughput is only 448 Mbps with 54 MHz transponder bandwidth. Thus the

transponder efficiency is 4.48 MbpsJ54 MHZ =0.08 bps/Hz. Each of these networks can accommodate

more than 40 VSATs in one transponder by assigning many VSATs to one SCPC carrier and operating

the group in the random access Aloha mode. The number of VSATs per group is determined by the

average duty cycle of their traffic. For each VSAT group the hub-to-remote link (Tables 2-3) can be

arranged in a broadcast time-dlvision-multtDlex (TOM) mode; that is, each station in the group

receives the same data stream from the bud station and then selec'_s the data destined for it, either

by searching for its own address or by se,ecting its own time slot in the TOM frame. The Aloha/1"DM

multiple access scheme can accommodate a very large number of bursty users but does not in

anyway improve the transponder efltmen,.-y (defined by the maximum number of simultaneous users)

which is very low. Furthermore, wl_en me 14/12-GHz space segment becomes crowded, the

interference into the adjacent satellites by remo(e VSATs and the interference into the remote VSAT

by adjacent satellites would be a malor DroOlem for VSAT networks using non spread spectrum

technology. However, at the present tame. tl_ese networks are indeed attractive, especially for the

single-hop architecture. For the full rnestt network where one remote VSAT communicates directly

with another remote VSAT, the non-sc)r, ad sl_ectrum technology using $CPC or Aloha as a multiple

access scheme would require a fre(2ue,_c'f syntr_eslzer that can operate over the entire transponder

bandwidth for full connectivity. Becaute o4 t_e _lgh cost of microwave frequency synthesizers, this

type of remote VSAT would cost more 'nan _'x_e used in the single-ho_ arch=tecture.

VI. VSAT Networks - An Ovenmm II1
Table 6.1. Network Parameters

Satellite

Transponder EIRP with 3.5 d8 Back-off

G/T

Hub Station

Antenna

Diameter

Gain (14 GHz)

Gain (12 GHz)

System Noise Temperature

G/T

HPA Output Power

Remote VSAT

Antenna

Diameter

Gain (14 GHz)

Gain (12 GHz)

System Noise Temperature

G/T

HPA Output Power

FEC Coding Gain @ 10-7 BER(Rate _/2)

PSK Bit Rate per VSAT

Number of VSATs

Tables 8.2-8.3

Table 6.4

36.5 dBW

+ 2 dS/K

5.5m

56.5 dB

55 d8

260 K

30.9 dB/K

8.5 dBW

1,8m

46.5 d8

45 dB

370 K

19.3 dB/K

2.3 dBW

4dB

56 kbos

40

80

V1.VSAT Network= -- An Overvie_ 12
Table 6.2. Link Analysis. Hub Statuion Transmits 40 Coded 112 kbps

Hub-to-Remote

Uplink

EIRP/Carrier (dBW) 49

Propagation and Pointing Loss (dB) 208.7

Satellite GIT (dB/K) 2

Boltzmann's Constant (dBW/K-Hz) -228.6

(C/1%). (dB-Hz) 70.9

Dowlink

ElRP/Carrier 17.5

Propagation and Pointing Loss (dB) 206.8

Station G/T (clB/K) 19.3

Boltzmann's Constant (dBW/K-Hz) -228,8

(C/N=), (dB-Hz) 58.6

Intermodulatlon and Interference C/_ (dS-Hz) 70.8

C/I_ (dB-Hz) 58.1

E_/No(dS) 10.8

Required F_/N+@ 10-t BER (dB) 8(FEC)

Margin for Rain and Implementation (dB) 2.8

Remote-to-Hub

48.8

208.5

2

-228.8

70.9

17.5

207

30.9

-228.8

70

70.8

85.7

18.2

12

8.2

VI. VSAT Network= .. An Overview

13
Table 6.3. Link Analysis. Hub.Station Transmit One Coded 4.48 Mbps TDM Carrier

Hub-to-Remote

Uplink

EIRP/Carrier (dBW) 85

Propagation and Pointing Loss (dB) 208.7

Satellite G/T (dB/K) 2

8oltzmann's Constant (dBW/K-Hz) -228.6

(C/R),, (dB-Hz) 86.9

Oownlink

EIRP/Carrier (dBW) 35

Propagation and Pointing Loss (dB) 206.8

Station G/T (dB/K) 19.3

Boltzmann's Constant (dBW/K-Hz) -228.8

(C/I_), (dB-Hz) 76.1

Intermodulation and Interference C/_ (dB-Hz) 83.5

C/t_ (dB-Hz) 75

F-e/N_(dB) 11.8

Required F,e/N_@ 10-_ BER (dB) 8(FEC)

Margin for Rain and Implementation (dB) 3.6

Remote-to-Hub

48.8

208.5

2

-228.8

70.9

15

207

30.9

-228.8

67.5

68

62.9

15.4

12

3.4

VI. VSAT Networks -- An Overview $ 4
Table 6.4. Link Analysis of a Full Mesh Network of 80 VSA Ts.

Each VSAT transrn=ts a Coded 112 kbps SCPC Carrier.

Uplink

EiRP/Carrier (dBW)

Propagation and Pointing Loss (dB)

Satellite G/T (dB/K)

Boltzmann's Constant (dBW/K-Hz)

(C/N_), (dB-Hz)

Oownlink

EIRP/Carrier (dBW)

Propagation and Pointing Loss (dB)

Station G/T (dB/K)

Boltzmann's Constant (dBW/K-Hz)

(CIN_), (dB-Hz)

Intermodulation and Interference C//. (dB-Hz)

C//_ (dB-Hz)

F-,IN,(de)

Required F.,JNe(_ 10-' BER (dB)

Margin for Rain and Implementation (dB)

48.8

208.5

2

-228.6

70.g

17.5

206.8

19.3

-228.8

58.6

70.6

58.1

10.8

8(FEC]

2.6

Vl. VSAT Networks - An Overview s s
ORIGINAL PAGE IS

OF POOR QUALITY

Spread Spectrum VSAT Networks

The use of $CPC as a multiple access scheme, especially in the full mesh network, is

constrained by frequency instability and phase noise. These constraints limit the data rate that can

be handled by the network. At a data rate of 56 kbps, the required frequency stability and phase noise

are 1 part in 10_ per 6 months and -85 dBc/Hz @ 10 kHz, respectively. To alleviate the precise

frequency assignment in SCPC, spread spectrum multiple access (SSMA) may be used.

The low power spectral density of spread spectrum signals allows the use of very small

antennas without significant interference to adjacent satellites, while the processing gain allows the

rejection of narrow band signals from adjacent satellites and terrestrial systems. In a SSMA satellite

system, each uplink earth terminal has its own addressed pseudo-noise (PN) code, and unlike the

situation in TDMA or FDMA where the carriers are separated by time and frequency, all active earth

terminals transmit in the same allocated bandwidth and overlap in time. Carrier separation is

achieved at the earth terminal by correlation of the received signal with the property addressed PN

code. Therefore, in a SSMA system each carrier in the group represents a low interference signal to

the others. The block diagram of a SSMA system is shown in Fig.28. To keep the system as simple

as possible, and to reduce equipment costs of the VSAT, all users transmit asynchronously.

Two main modulation techniques are considered: phase shif_ keying (PSK/QPSK) and wide band

M-ary frequency shill keying (MFSK). The precise performance of an asynchronous PSK-SSMA

system is difficult to compute because an evaluation of the partial cross-correlations between PN

codes is required [3]. When the user population is large, and the PN codes have low mutual

cross-correlations, the ensemble average performance is normally considered knowing that some

combination of users may have worse _erformance than the ensemble average performance and

some a better performance. The ensemble average performance may be derived as follows. Let us

assume equal carrier power C for each _$AT The inter-user interference Dower spectral density is

kC/P_ =. kF-tP_/R==. kE_/N where k-,- I ,s ',he number of simultaneous users in the system, _ is the bit

rate, R_ is the PN chip rate, N = ,Q.,_, s '_ne PN code period, and F_ is the bit energy. Therefore the

VI. VSAT Networks - An Overview 8S
_o_e and

l_ter_ere_ce

PN Code ?N Code

Figure 28. SSMA Modem

VL VAST Network--An Overview 17
ORIGI/_L PAGE iS

OF POOR QUALITy,

total noise power spectral density is k_/N + t_ where N_is the thermal noise power spectral density

Consequently, the total energy per bit to noise density ratio is

Eo 1

kEojN + NO k/N + (E_/NO)-1

(e.1)

and hence the PSPUQPSK-$$MA bit error probability is

f

P_ = Q('V'2/[k/N + (_/NO)-_]) (6.2)

where Q(x) is the Gaussian integral defined as

O(x) = 1 fx _ -f'/=Oy -_- e (6,3)

,J2_

As an example, consider a single-hop QPSK-SSMA VSAT network with the same network

parameters shown in TableS.1. The hub-to remote link is a coded 3 92 Mbps TDM carrier, and there

are 35 remote-to-hub links with QPSK-SSMA carriers, each with a data rate of 58 kbps. The PN code

period is 1023 and the QPSK-SSMA signal _s filtered to 54 MHz bandwtdth (T=B == 0.8). A link analysis

similar to Table 6.3 shows that _/N_ = 158 dB in clear air. With k = 34, the link uncoded bit error

probability is 10-t with no margin for ram attenuation and implementation for remote-to-hub links.

As a comparison, the SCPC-TDM system _n Table 6.3 allows 40 carriers at 56 kbps carriers for the

remote-to-hub links and a 3.4 dB margin As a second example, we consider a full mesh QPSK-SSMA

VSAT network with the same network parameter and a total of 80 QPSK-SSMA carriers. A link

analysis similar to that In Table 6.4 shows that E_//_ = 10.6 dB in clear air. With k = 79 the link

coded bit error probability is 10-_ w,th 1 4 dB margin.

In the above two examples the SCPC systems outperform the SSMA systems, but the difference

is small. The advantage of the SSMA system _s that no frequency synthesizer is required, which could

significantly reduce the cost of the 'u_= mesh VSAT network. In addition the lower power spectral

density and the interference relect'on cal3alo_l_ty of SSMA systems make it attractive for 6/4..GHz

VI. VSAT Networks - An Overview SI
operation. It is doubtful that the interference levels generated by VSATs using SCPC transmission

could be tolerated with 2= satellite spacing unless a large antenna (5 m or more) is used.

When wideband MFSK is used as the modulation scheme, the performance of VSAT networks

can be improved. The performance of an MFSK-SSMA system is shown in Table 6.5. It is seen that

MFSK-$$MA can provide a much larger number of simultaneous users than PSK/QPSK-SSMA. For

example, at 10"_ BER and link _JN= = 15 dB, MFSK-SSMA can accommodate 70 users for M = 16 while

QPSK.SSMA can accommodate only 46 users. A single hop MFSK-SSMA (M =16) VSAT network with

the same network parameters in Table 6.1 would require a link _lN_ = 12 dB to achieve a 10-T BER

with 40 remote VSATs. This would yield a margin of 3.4 dB. A full mesh MFSK-SSMA (M = 16) VSAT

network with the same network parameters and a total of 80 MFSK-$SMA carriers achieves a coded

bit error probability of 10"_ with a margin of 2 dB. A summary of the performance of these multiple

access systems is given in Table 8.8.

VL VSAT Networks - An Overview II9
Table 6.5. Performance of MFSK,SSMA @ _o-' 8ER

-" 56 kbpS, B = 54 MHz, Code Length = 511.

F_jNo(dB) M 3/'(MHZ) Users (k + 1)

16 16 2.8 80

15 16 2.8 70

14 16 2.6 60

9O

VI. VAST Network--An Overview
Table 6.6. Summary of VSAT Systems

Architecture

SCPC/SCPC single hop

SCPC/TDM single hop

$CPC full mesh

MFSK/DS full mesh

MFSK/DS single hop

QPSK/DS single hog

QPSK/D$ full mesh

Users System Margin Throughput

40 6.2/2.6 8

3.4/3.6 8

80 2.6 8

80 2.0 8

40 3.4/3.6 8

35 0.0/3.6 7.2

80 1.4 8

oO

V1.VSAT Networks - An Overview 11
Epilogue

ORIGINAL PAGE IS

OF POOR QUALITY

We have presented an overview of VSAT networks, a new bypass technology. Twenty-five

precent of the large corporations in the United States bypass the local and long-distance telephone

systems and the trend has been accelerating s_nce the breakup of the Bell System in 1984 under a

court-supervised anti-trust agreement untJI '.he nation implements a wholly digital transmission

network called the Integrated Services 3,g_taJ Network. (ISDN), bypass systems will continue to grow.

and this undoubtedly will favor develooment of more low-cost VSAT networks.

The technology for VSAT networX= ,s _ere It _s just a matter of selecting the right architecture

and the right multiple access scheme for ',_e intended service. Because VSAT networks are severely

power-limited, it makes little difference to _n_ performance whether non-spread spectrum or spread

spectrum technology is used. However '_e ,nterference received at adjacent satellites is much lower

when a VSAT uses spread spectrum _<:x:u_at=on Interference control in VSAT networks can also be

improved by using rectangular antenna rejectors By orienting the :nger of the antenna dimensions

in the plane of the geostationary arc. tree adlaCent satellite interfere_-e can be reduced while keeping

the overall antenna dimensions reaso_aOhf compact. Non-_pread speclrum systems such as

SCPC-TDM are favorable for the smgle-.e, oO arcn,tecture operating at 14/12-GHz band, while in this

band QPSK-$SMA or MFSK-SSMA m_r_ _ preferable for the full mesh architecture. The average

transponder efficiency in the 14/12-GHz ooerat,on ,s expected to be around 008 bpslHz for either

technology. For 6/4-GHz operation the .o_ :_wer spectral dens=ty and II_e =nterference rejection

capability of spread spectrum systeml ,'_a,,e ,hem ;_referable for VSAT networks. The transponder

efficiency in the 8/4-GHz operation s :e_,_dent rot only on the power constraints of VSAT networks

but also on the adjacent satellite rTe,"e,t,,¢e and ,errestrlal m=crowave interference. The estimated

efficiency in this band is about 0 (_3:0t. -z

The future of VSAT techno_<:_, _ O_om,slng. at least for another decade betore ISDN

becomes available at all locations , t e..e* :,o_,1 Most VSAT networv, s will be implemented in the

14/12-GHz band to take advantage :_ '"e., _e,e¢ antenna gain and the relatively free interference

VI. VSAT Networks -- An Ovtrwew g2
environment. Thus they can provide higher data rate services for both data and voice

communications for a larger number of widely distributed users at much lower cost than that of

terrestrial telephone systems. For VSAT networks to use the satellite resources (i.e. bandwidth)

efficiently, a new generation of VSAT satellites must be designed to take into account the severe

power constraints imposed on VSAT networks. Recall from the link design shown in Tables 6.2- 6.4

that a typical 14112-GHz VSAT network transmtts only 4.48 Mbps in a 54 MHZ transponder. For a

non-spread spectrum system, a 10 MHz transponder would suffice. This represents a bandwidth

saving factor of 5.4. A typical domestic 14/12-GHz satellite with dual polarizations has 16 54

MHz.transponders spanning a total bandwidth of 500 MHZ. A satellite designed for VSAT operation

could have 16 10 MHz-transponders spanning a total bandwidth of only 93 MHz (or 80 10 MHz

transponders spanning 500 MHz). If growth in VSAT networks proves to be as explosive as it is at

present, there may be a demand for more 14/12-GHZ satellites in the geostattonary arc. By designing

VSAT satellites especially for small terminals, the geostationary arc would be able to accommodate

more satellites to meet business demand, and the capital cost of the networks could be lowered.

Vl. VSAT Networks - An Overview 9 )
ORIGINALPAGEIS

OF POORQUALITY

VII. Conclusions

The analysis analysis in the previous chapters demonstrates that spread spectrum multiple access is

feasible, though still inefficient relative to non spread spectrum systems. The MFSKJDS systems

proposed can achieve bandwidth efficiencies up to 11% for the Ku band example (58 kbps) and for the

C band example (1200 bps). When the power and interference :onstraints of practical satellite

systems are considered, the throughput is reduced to approximately _ for the Ku band example and

3% for the C band example. The advantage of spread spectrum sTstems is that they make VSAT

networks possible when operated at C band and when the angular satellite spacing is small, situations

Which make VSAT networks infeasible for other modulations. The construction of SSMA VSAT

networks, because of their inefficient use of satellite bandwidth, is mainly a question of economics.

If transponder leasing costs remain low, SSMA VSAT networks may proliferate. Presently, networks

exist which are far less bandwidth efficient than the systems presented here but are still commercially

successful [33].

Spread spectrum overlay will probably be tmplemented by a small number of satellite users that need

an extra low data rate channel. The results of Chapter 4 predict that OS overlay signals with data

rates of approximately 3000 bps can be superimposed on general satellite traffic. When convolutional

coding is used, the data rate may be ,ncreased to 10 to 20 kbgs. While these data rates are generally

insufficient to accomodate even a s_ng[e ,.'o_ce channel, they can accomodate several low rate

VII. Conclusions g4
terminals. Because of the overlay Systems' limited performance, their commercial appeal will

probably be limited.

On.board processing has the ability to dramatically increase the throughput of SSMA systems, as

seen in the near doubting of the throughput Ln our example. The Ku band MFSKJD$ multiple access

system considered in Chapter 3 could accomodate 57 simultaneous users. The addition of a

processing transponder which converts the MFSKJDS uplink to a TDM downlink increases the number

of simultaneous users to 110. The future of on-board processing in the commercial market, however,

is probably not bright. The complex on-Ooard electronics required would make the satellite very

expensive, and would probably decrease Lts rehal:)llity. More importantly, the transponder becomes

dedicated to a certain type of servtce. The al:)llity of the classical transponder to handle general

signals allows it to be leased to many dtff'erent users at different times. This makes its commercial

success independent of the success of an individual company or specific system, an advantage that

the dedicated processing transponders do no( enloY.

Vll. Conctu=lone gS
IIX. References

[1] Timothy Pratt and Charles W, Bostian, Satellite Communications, New York: Wiley, 1988 p. 225.

[2] S. Josel=h Campanella, and John V. Hardngton, "Satellite Communications Networks °,

Proceedings of the IEEE, Novemi0er 1984, p. 1507.

[3] T,T. Ha, Digital Satellite Communications New York: Macmillan, 1988. pp. 206-213.

[4] Pratt and Bostlan, p. 235.

[5] Campanella and Harrington, p 1510.

[8] Robert C. Dixon, Spread Spectrum Systems. New York: Wiley, 1984, pp. 15-28.

[7] Dixon, pp. 28-42.

[8] Robert M. Gagliardt, Satellite Communications

Publications, 1984, pp. 267-268.

[9] T.T. Ha, pp. 538-539.

[10] T.T. Ha, pp. 539-54,4.

Belmont, California: Lifetime Learning

[11] T.T. Ha, p. 543.

[12] T.T. Ha, pp. 544-548.

[13] MB. Pursley and D.V. Sarwate "Performance Evaluation for Phase-Coded Sl_read Spectrum

Multiple Access Communication I_ Code Sequence Analysis," IEEE Transactions on

Communications, Vol, COM-25, No 8, August 1977, pp. 800-803.

[14] Gagliardt, pp. 282-288.

[15] Rodger E. Zlemer and Roger L Paterson, Digital Communications and Spread Spectrum

Systems, New York: Macmttlan '_85 ;_1_569581,585

9e

IIX. References
[lS] y. Yamauchi, M. Sato, N. Morinaga. and T Namekawa. "Improvement in Multiple Access

Capability in M-ary-FSK/Direct Sequence SSMA." EtectronLcs and Communications in Japan. Part

I, Vol. 69, No. 3, pp. 86-91, 1986

[17] Pratt and Bostian, p. 274.

[18] Andrew J. Viterbi, "A Processing Satellite Transponder for Multiple Access by Low-Rate Mobile

Users," Proceedings of the Fourth lnternat=onat Conference on Digital Satellite Communications,

Montreal, Canada, Octol0er, 1978. pp 166-174

[t9] T.T. Ha, pp. 560-561.

[20] Pratt and Bostian, p. 250.

[21] H.J. Kochevar, "Spread Spectrum Multiple Access Communications Experiment Through a

Satellite,* IEEE Transactions on Commumcat=ons, Vol. COM-25, No. 8, August 1977, p. 855.

[22]

[23]

[24]

Yamauchi, pp. 86-91.

T,T. Ha, pp 544-548.

K. Yap, "Error Probability of Asynchronous Spread Spectrum Multiple Access Communication

Systems," IEEE Transactions on Commumcations, Vol. COM-25, No. 8, pP. 803-809, August 1977.

[25] George C. Clark, and J. Bibb Cam. Error Correction Coding for Digital Communication, New YorJc

Plenum Press, 1982, p. 369.

[28] A. Pal_0ulls, "Narrow Bend Systems and Gaussianity', IEEE Transactions on Information Theory,

January, 1972.

[27] V.K. Prabhu, "Error Rate Considerations for Coherent Phase-Shift Keyed Systems with

Co-Channel Interference," The Bell System Techinical Journal, ._larch 1969, pp.743-767.

[28] T.T. Ha, p 202.

[29] T.T. He, p. 538.

[30] T.T. Ha, pp. 150.

[31] T.T. Ha, pp. 561.

[32] Pratt and Bostlan, p. 273.

[33] E.B. Parker, "Micro Earth Stations as Personal Computer Accessories," Proceedings of the IEEE,

vol. 72, November 1984, pp. _526-_531

[34] T.T. Ha, pp. 428.433.

[35] E. B. Parker, "Cost-effective 0ate Communications for Personal Computer Applications using

Micro Earth Stations, "IEEE Journal o_ Selected Areas in Commumcatlons, vol. SAC-3, No. 3, pp.

449-456, May 1985.

[36] A. Viterbi, "When Not to Sl_reed 5Oec"trum-A Sequel'. IEEE Commumcations Magazine, vol. 23,

pp. 12-17, Apr. 1985.

IIX. References ORIGINAL PAGE IS

OF POOR QUALITY

g?
[37] D. Chakrabortz, "Constraints in Ku-band Continental Satellite Network Design", IEEE

Communications Magazine, vol. 24. no. 8, pP. 33-43, August 1986.
Appendix A. Performance of Noncoherent MFSK in

Colored Noise

An MFSK modulator transmits M distinct symbols

S,,S=..... su

A noncoherent MFSK detector is composed of M parallel channels each containing a matched filter

for one of the M symbols (Figure 5). The sampled outputs of these filters define a set of observation

random variables

r,r_...., r_

The probability that symbol q is chosen ,n the receiver given symbol p was transmitted Is denoted

by

pr (_ > r.Is,)

Appendix A. Pirformance of Noncoherent MFSK in Colorld Nolse

II
Intheusualcasewhere the interference is additive white Gaussian noise and the spacing between

symbols satisfies the orthogonality condition the output of a channel whose symbol has been

transmitted is the envelope of a signal plus noise, and is described by a Rician probability density

function. The outputs of the other channels are the envelopes of Gaussian noise, and are described

by Rayleigh probability density functions.

For the case where the noise is GaussJan but colored and has a PSD described by the function

e(f) the appropriate probability dens=ty functions are [34]

p(r, ls,) = _2+ex=f

;z + E_

E,N,

(Rician) (A 1)

pGIsp) == I

_--_""_PL- E_,j

(Rayleigh) (A2)

where N_ =. _(_) , N+ = ¢(_)

and E, is the energy per symbol .

/, ,s the moclifled Bessel function of order zero,

Pr(to> r,is,)=

',o<r,'_) / "pCr_ISo)dr_dr.

| D

I"_+r s.) exp IEJ_I= dr.

Appendix A. Performance of Nonco_,ent _FSK m Colert_l Notml 10 0
E,___.____

_+N_

Now making the substitution

, r,'N,

Z =m

I _'÷el I

/% {'" 2z exD - I.

J-4

2z dz

Multiplying by

inside the integral and compensating o_Isa(le tl_e integral gives:

-- z'+ ' N,+ N,

"0

llol 2Z

r_,

dz

now let

Appendix A. Performance of Nonc:_ Ulr_K m Colored Noise 101
ORIGINAL PAGE IS

OF POOR QUALITY

and

z E,,%

2

and the expression becomes

N, exp _' _T ex_ J.

_.+"-'E '_.° _ _0 _ 2o'

dZ

Since the function inside the integral ,s • _¢_an pdf, the integral = 1. gJ_,=ng the final result:

Pr(r_>r, %) N', expl E, 1 (A3) - ----_-_ N_-+-'._

When the noise power spectral densJl_ .t _at N, =. N,_ and this equat=on _s _dentical to that for the

probability of error for noncoherent o_t_•_ _:SK ,n ,_mte noise.

Appendix A. Performance of No_q:=_ _II_ = CmrqKl Noise I 0 2
Appendix B. Derivation of Interference PSD

The power spectral density (PSD) of a pseudo-noise (PN) sequence is of the form

sin(af) af ] = (B1)

• The envelope of this function can be approxtmated by

(S2)

1 4-aZf 2

The power spectral density I(f) of an FSK tone spread twice by being multiplied by two uncorrelated

PN codes can therefore be approx=mated by a convolution of the approximating function above wltl_

itself

. 1 (83)

-,- a zfz 1 +. aZf z

Appendix B. Derivation of Interference PSO 10 3
I(f) = 1

1 + aZk z

-oo

1

X

1 + a z(f_ k)Z

dk

= 1

== . X

1 + aZk z

k Zaz -2a ztk 4- fZa z+ 1

dk (B4_

I" f

Ik+n

= ---- dk ÷

1 + aZk z

mk +p

kZaZ -2aZkf+ fZa Z+ 1

dk

where I, n, m, and p are partial fraction expansion coefficients which satisfy

(/k + n)(kZa z -2a zkf+ fZaZ +1) + (ink +p)(1 + aZk z) =,

(ss)

Equating like coefficients of powers of k and solving gives

3

p --

4 4- a zfz

m =z

-2

x (4 + a zfz)

2

/ =

x (4 + a zfz)

1

/3 ==

4 + a zfz

Appendix B. Derivation of Inter/erenc= PSO 1 0 4
EVALUATE INTEGRALS:

=" 1 Ik+n + aZk z

dk =

f _" 1 -+-/ka2k z

dk -t-

== 13 1 +aZk z

dk

n

/3'it

Iii i

8

mk dk

k za a _ 2a Zkf + fza z + 1

Appendix B. Derhtstlon of Interferen<e P$O 105
= mk -- dk

kZa z-2azkf+ fZaZ + 1

+

I= kZaZ- 2aZkfp + fZa Z+ 1

dk

m In(aZk z 2aZkf+aZf z+l) l =,

,,_ z - -=,

-'d

-tmx _ a Zkz -2a Zkf dk + fZa Z+ 1

+

I= ark =- 2aZkf+p fZaZ + 1

dk

= 0

m_x P_

a a

Combining terms gives

n_

/(f) = T

m_f

a

p_

a

(Bs)

Substituting the values for n, m, and 0 gwes

I( f) ==

2rt

a

4 + a zfz

(BT)

Normalizing the function to have an area of 1 by multil_lying by a constant c

Appendix B. Derivation of Interference ?SO loe
= c

2_r

= 1

2a a

a z

C _ m

z

which gives a normalized interference PSO function of

2a

_(f) =

_(4 4- a Zfz)

(aS)

For a PN code with a code rate of R=. a ,=

Re

Substituting this into equation (B8) alxTve gives a normalized interference PSi3 function of

I(f) -

2R_

4R_ 4- 2fz

(B9)

Appendix B. Derivation of Interference _10 I 0 7
ORIGINAL PAGE IS

OF POOR QUALITY

Appendix C. MFSKJDS/FDM Performance Analysis

Program

C

C

C

C

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C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

THIS PROGRAM EVALUATES THE BER FOR A DS/MFSK MULTIPLE ACCESS

SYSTEM. THE PROGRAM CALCULATES THE ALLOWABLE NUMBER OF USERS

IN A MFSK/DS SSMA SYSTEM FOR A GIVEN SYSTEM BIT ERROR RATE AND

VALUES OF EB/N0 FROM 10 TO 18 DE}.

INPUT PARAMETERS:

RC - THE CODE (OR CHIP RATE) OF THE SPREADING COOE

IN BPS

DELF - THE FREQUENCY SPACING BETWEEN MFSK FREQUENCIES

IN HZ

M - THE NUMBER OF MFSK FREQUENCIES

K - LOG BASE TWO OF M

TB - THE BIT DURATION OF THE DATA SIGNAL = 1/RB

BER - THE DESIRED SYSTEM BIT ERROR RATE

NO - THE NOISE POWER SPECTRAL DENSITY (ALWAYS NORMALIZED

TO 1)

OUTPUT PARAMETERS:

N - THE NUMBER OF USERS

PROGRAM OPERATION:

AVALUEOFEB/N0(=10) _SSELECTED THE BERIS CALCULATED

FOR INCREASING VALUES C; N LjNTIL THE BER IS GREATER THAN OR

EQUAL TO THE DESIRED S_STE_I BER THIS VALUE OF N IS THEN

OUTPUT AND THE PROCESS _N _EPEATED FOR THE NEXT EB/N0.

APPLICATION TO MFSK/DS, FOM SYSTEM PERFORMANCE

THIS PROGRAM IS USED ?0 CALCULATE THE NUMBER OF USERS THAT

AppendixC. MFSK/DSIFOMPlrlorm=ncmAnary=,=Program 1011
C

C

C

C

C

C

CAN BEACCOMODATED IN AN FOM SLOT. TO FIND THE TOTAL NUMBER

OF SYSTEM USERS, THIS PROGRAM'S OUTPUT MUST BE MULTIPLIED

BY Z, THE NUMBER OF FDM SLOTS.

COMMON RC,C,DELF,N,M,K,TB,N0

RC = 210000.00

DELF = 308000.00

M;18

K=4

TB -- 1/56000.0

BER = 1.E-08

N0"-I

DO 200 J=&10

C = 10."((J-1)/&)frB

DO 100 N=1,500

Y=E(N)

IF (Y.GE.BER) THEN

WRITE(&') C,N

GO TO 2OO

ENOIF

100 CONTINUE

200 CONTINUE

STOP

ENO

100

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

FUNCTION PHI(I)

FUNCTION TO EVALUATE THE INTERFERENCE I:_WER FUCTION

EVALUATE8 EQUATION NUMBER 2.12 IN CHAPTER 2

COMMON RC, C,DELF,N,M,K,TB,NO

PI = 3.141592654

PHI =0.0

DO 100 J = 1,M

PHI = PHI + 1.0/(4.'RC'"2 + (Pl'(l-J) "DELF)"2)

CONTINUE

PHI = 2.*RC'C*(N-1)/M"PHI + NO

RETURN

END

FUNCTION E(N)

EVALUATES THE BER OF T_E SSMA SYSTEM AS A FUNCT1ON OF THE

NUMBER OF USERS IN THE SYSTEM USING EQUATION 2.9

Appendix C. MFSI(JOS/FOM performe_e Ana_sl= Pt.ogram 109
C

C

C

C

C

100

2OO

C

REAL't6 VI,V2,PROD,SUMONE

FUNCTION EVALUATES THE PRO8 OF BIT ERROR FOR DS/MFSK

INTEGER P,Q

COMMON RC,C,DELF,N1,M,K,TB.N0

SUM =0.0

DO 200 P=I,M

PROD = 1.0

TP = PHI(P)

ONE = 1.0

DO 100Q=l,M

IF(Q.EQ.P) GOTO 100

TQ = PHI(Q)

VO-'- -C'TB'K/(TP + TQ)

Vl = EXP(V0)

V2 = ONE-TQ/(TQ + TP) "V I

PROD = PROD'V2

CONTINUE

SUM = SUM + (1. D0-PROD)

CONTINUE

E = SUM/(2.*(M-1))

RETURN

END

Appendix C. MFSKJDSIFDM PIrform_ll A,_lrl, ln I_plm 1 1 0
Appendix D. MFSK/DS/FH Performance Analysis

Program

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

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C

10

C

THIS PROGRAM EVALUATES THE BER FOR A FH/DS/MFSK MULTIPLE ACCESS

SYSTEM.

THE INPUT PARAMETERS ARE INOENT]CAL TO THOSE OF THE MFSKJDS

PROGRAM AND ARE READ FROM UNIT NUMBER 4.

Z IS THE NUMBER OF FH SLOTS }N THE SYSTEM BANDWIDTH

NMIN - THE START VALUE OF THE NUMBER OF USERS IN THE ITERATIVE

CALCULATION OF THE NUMBER OF USERS

INC - THE iNCREMENT APPLIED TO THE NUMBER OF USERS

PROGRAM OPERATION:

THE PROGRAM STARTS WITH THE NUMBER OF USERS N = NMIN,

IT THEN COMPUTES THE BER FOR THIS N. IF THE BER IS LESS

THAN THE DESIRED SYSTEM BER THEN N IS INCREMENTED BY INC

AND THE CALCULATION IS I:_RFOI_MED AGAIN, UNTIL THE BER IS

GREATER THAN OR EQUAL TO T_E DESIRED BER. THIS VALUE OF N

IS OUTPUT AND THE PROCESS 'S _EPEATED FOR A NEW EB/N0.

REAL NO

COMMON RC,C,DELF,M,K,TB

READ(4,10) RC,C,DELF,M.K.RB _:) 8E =t Z

FORMAT(F20.5,/,F20.5,/,F20 5, ,_3 ,'_3, ;20 5J.F20.5,

$ /,F20.SJ,F20.5)

BER = 1.E_8

WRITE(7,12) RC, DELF,M,K.I_8 _(_ EIE_ Z

Appendix O. MFSKJDS/FH Perlorm=_¢= &_eM _ 1 1 1
12

C

C

100

C

C

C

FORMAT(lX,'HYBRID FH/DS/MFSK SSMA PERFORMANCE',//,3X,

$ 'CODE RATE = ',F15.2,/,3X,

$ 'DELTA F " ',F15.2,/,3X,

$ 'M - ',16,/,3X,

$ 'K = ',16,/,3X,

$ 'RB = ',F15.2,/,3X,

$ 'NO = ',F15.2,/,3X,

$ 'BER = ',E15.2,/,3 X ,

$ 'FH SLOTS = ',F15.2,//, 6X,'EB/N0 _SX _ /

TB = I/RB

INC=2

C

200

300

C

C

C

C

C

C

C

C

C

C

C

NMIN=5

DO 300 J= 8,10

EBNO = (J-1)'2.

C = 10.'*((J-1)/5.)/TB

DO 200 N = NMIN,5OOO,INC

PBE =0.0

DO 100 I=I,N

PBE = PBE + W(N, I,L::q

CONTINUE

IF(N.GE.100) INC = 15

IF(N.GE.250) INC = 75

IF(N.GE.1000) INC = 200

IF(PBE.GE.BER) THEN

WRITE(7,') EBNO,N

NMIN=N

GOTO 300

ENDIF

CONTINUE

CONTINUE

STOP

END

100

C

FUNCTION PHI(I,N)

FUNCTION TO EVALUATE "ruE h,'IE_¢E#ENCE POWER FUCTICN EQ 2.12

REAL NO

COMMON RC,C,DELF,M,K,T8 _13

PI =3.141592654

PHI =0.0

DO 100J=l,M

PHI = PHI + 1.0/(4.'RC"2 - _'"

CONTINUE

".E.=,"2)

PHI = 2.'RC°C'(N-1)/M "pHI " ._O

Appendix O. MFSI(JDS/FH Perfor,,q_= _" _ _ _ 2
C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

C

RETURN

END

tO0

200

C

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C

C

C

C

C

C

C

C

C

C

C

FUNCTION E(N)

EVALUATES THE MFSK/DS BER AS A FUNCTION OF THE NUMBER OF

USERS

REAL'16 VI,V2,PROD,SUM,ONE

REAL NO

FUNCTION EVALUATES THE PROB OF BIT ERROR FOR DS/MFSK

INTEGER P,Q

COMMON RC,C, DELF,M,K,TB,NO

SUM =0.0

ONE = 1.0

DO 200 P=I,M

PROD = 1.0

TP= PHI(P,N)

DO 100 Q = 1,M

IF(Q.EQ.P) GOTO 100

TQ = PHI(Q,N)

VO=-C'TB'KJ(TP + TQ)

Vl = EXP(V0)

V2 = ONE-TQ/(TQ + TP_"V1

PROD = PROD"V2

CONTINUE

SUM =SUM +(ONE-PROD)

CONTINUE

E= SUM/(2.'(M-1))

RETURN

END

FUNCTION COMB(N,K)

FINDS THE NUMBER OF CCMB_NAT'CNS OF N OBJECTS TAKEN K AT A TIME

Appendix O. MFSKJDS/FH perfotmsn<e Anel,l=,s Program

113
C

C

C

C

tO0

C

C

C

C

C

C

C

C

C

C

C

REAL'16 PROD

INTEGER DENOM

IF(N-K.GT.K) THEN

JMIN =N-K+1

DENOM = K + 1

ELSE

jMIN =K-t-1

DENOM = N-K + 1

ENDIF

PROD = 1.0

DO 100 J=JMIN,N

DENOM = DENOM-1

PROD = PROD*J/DENOM

CONTINUE

COMB = PROD

RETURN

END

FUNCTION W(N,K,Z)

EVALUATES THE BERNOULLI TRIALS FORMULA

COMB(N,K)'P"K*Q"(N-K) WHERE P AND Q ARE DEFINED BELOW

USES LOGARITHMS IF THE VALUES OF THIS FUNCTION _RE

SUFFICIENTLY SMALL TO CAUSE UNDERFLOW IN THE COMPUTER

P = 1./Z

Q=I.-1./Z

IF (N.LE.200) THEN

R =ALOGIO(COMB(N.KI)

ELSE

DO= FACLOG(N)

IF(K.LE.50) THEN

D1 - ALOGIO(FACT(K))

ELSE

D1 = FACLOG(K)

ENDIF

IF(N-K.LE.50) THEN

D2 = ALOG 10(FACT(N-K))

ELSE

02 - FACLO G(N-K)

ENDIF

R = 00-O1-O2

ENDIF

X=E(K)

IF(X.LE.0.0) THEN

U =-50

ELSE

U =ALOGIO(X)

Appendix D. MFSKJDS/FH Pllrformlr_:e Anatysl= Program i _ 4
C

C

C

C

C

10

C

C

C

C

C

C

C

ENDIF

W=U + K'ALOG10(P) +(N.K)'AL.QG10(Q) 4-R

IF(W,LT.-60.) THEN

W=0.0

RETURN

ELSE

W = 10.'_/

RETURN

ENOIF

END

FUNCTION FACT(K)

COMPUTES K FACTORIAL

FACT - 1.

DO 10 J--2,K

FACT- FACT'J

CONTINUE

RETURN

END

FUNCTION FACLOG(K)

THIS FUNCTION COMPUTES THE COMMON LOG OF

STIRLING'S APPROXIMATION TO K

P! = 3. i415926538

E = 2,7182818284

FACLOG = K'ALOG10(K/E) + 0 5"A LOG 10(2.'PI'K)

RETURN

END

Appendix D, MFSKJDS/FH Perfo_ _ _ I 1 s
Part B. Chirp Spread Spectrum System and

Overlay Service

Part B. Chirp Spread Spectrum System and Overlay Service
ORIGINAL PAGE IS

OF POOR QUALITY

1. Introduction

The pulsed FM or chirp modulat=on technique, invented by B. M. Oliver at Bell Labs in

1951 and formulated by C. E Cook in 1958, has been widely used in radar systems as a

solution for the conflicting requirements of simultaneous long.range measurements and high

range resolution. [1]. The range resolution is obtained by transmitting a long frequency

modulated pulse that is time coml_ressed in the receiver _nto a much narrower pulse. The

energy distributed in the long pulse _s coherently processed in a matched filter receiver to

yield a short pulse of enhanced aml_lltude. [1], [2] A detailed account of the chirp concept and

the relevant mathematics has s=nce been given by many authors [1], [21, [3], [4], [5]

The physical constraint of budding delay lines that have dispersion characteristics which

provide the required high Time-BanawtOth product have been overcome recently by using

surface acoustic wave (SAW3 devices and other types of dispersive delay lines (DOL). Chirp

has now become an effectwe Ioo_ for producing small, low cost, lightweight spread spectrum

systems for communication and _angmg systems, as well as for radar. [1]

Spread spectrum tecnntQues Rave made the use of small earth terminal satellite

networks commercially valuaOle k:_ _ow data rate users. The low power spectral density of

spread spectrum signals al_o_,_ t_e use of very small antennas w=thout significant interference

to adjacent satellites w_e_ _ansm,rlmg. whale the processing gain allows the rejection of

1. Introduction I i T
ORIGINAL PAGE IS

OF POOR QUALITY

signals received from adjacent satellite when receiving. Among the four basic types of spread

spectrum techniques, chirp spread spectrum systems have been proven to have advantages

over other techniques tike Direct Sequence (DS), Frequency Hopping (FH), Time Hopping (THt,

and various hybrids. No coding is necessary, in principle, and thus no code synchronization

is necessary either, In addition, multiple access can be achieved with simple sweep rate

changes rather than by using different pseudo random sequence codes. [6], [7], The low

vulnerability of chirp systems to deliberate jamming has been the major obstacle to their

application for military use. Thus, the technique has received little attention.

In a chirp system, all non-chirp type interfering signals and noise at the input of the pulse

compression filter appear at the output either with the same mean power or with decreased

power. However, the desired chirp signal gets enhanced by the processing gain, resulting in

the rejection of the interference and noise. The possibility of relatively simple equipment

implementation makes the cmrp s0read spectrum system attractive for low data rate, bursty

commercial communication systems.

The feature which makes cmrp attractive for low data -ate satellite communications is

the absence of a synchronization loop in the receiver, In all other spread spectrum systems,

the receiver must synchronize to the PN sequence used to spread the original data. If data

rates are low and the messages are snort, the receiver may take many seconds to

synchronize to the received s;gnal. Lnorder to receive a message less than a second in length.

The lower the data rate and the longer the PN sequence, the more pronounced this problem

becomes, and very low througnl_ut qs achieved. A bursty system, in which messages are

transmitted infrequently to a g=ven recewer, makes the problem even worse, because receiver

synchronization cannot be maintained between transmissions. It is in this application that the

chirp system has the best _er_crmance. since no receiver synchronization is needed, and

throughput can be maintained at '0G°'o

In this report, severa= :_,rp spread spectrum systems are studied, using different

modulation techniques. T_ese _ctude within chirp, and multiple chirp, where single or

multiple chirps modulate the :3rr_er 3mpl_tude, freduency or phase. Based on the preliminary

I. Introduction I _ I
study, the proposed Coded Multiple Chirp Spread Spectrum System is described in detail,

together with a performance analysis. This forms the maior part of the work in Task 2. In

addition, the possibility of Multiple Access (SSMA) using the proposed system is also

evaluated as the final step in Task 2.

For Task 3, the overlay system study, interference analyses were performed when the

proposed chirp spread spectrum system is overiayed on an existing FM-TV satellite link in

C-band and Ku-band. In this study, various models of FM-TV signals are used to find a general

expression for interference which can be used to find the error signal in an FM-TV

discriminator. Finally, typical FM-TV signals are modeled with a linear multi-step function and

a random step function, and are then overtayed with the chirp signal. By reviewing the

spectral and time domain analysis, an optimum overlay method is proposed based on the

carrier to interference ratio. The best offset frequency for the chirp signal and some

performance evaluation data which support this method are presented.

1. Introduction 1 19
ORIGINAL PAGE IS

OF POOR QUALITY

2. Review of Chirp Spread Spectrum Systems

Because of the inherent capability of interference rejection, the chirp modulation

technique is a strong canmdate for use in spread spectrum communication systems. A

number of authors have proposed cmrl= signals for the representation of binary digital

communication by assigning an _ncreas=ng frequency expanded pulse known as 'up-chirp' to

a binary '1' and decreasing frequency expanded pulse known as 'down-chirp" to a binary '0"

(or vice-versa). [2], [7], [8] The aegree of orthogonality between these two signals determines

the possibility of overlapping t_me-rrequency regions, and thus the efficiency of bandwidth

utilization. Berni and Greg [9] compared BPSK. chirp and FSK and found that chirp signalling

performance is between the performance of BPSK and FSK from the power and bandwidth

utilization viewpoints [2]. Hlrt and Pasupathy [10] investigated the effect of designing chirp

signals with continuous phase at 0,t trans=t_ons, but the use of these techniques increases the

receiver complexity.

The use of up and clown cmrD s,gnal for the spread spectrum system was suggested in

1962 by Winkler [7] Daylon ,,1 'ound that these signals gave excellent performance in

multipath situations due _o _e _arrow compressed pulse width. In addition, various

applications such as h,q_.tre_uency clata transmission [12] and air-ground satellite

communication [13] have _n ,e_e_ [14] Recently Baier et al. [15] had suggested the

2. Review of Chirp Spreul So4J(u'w_ Syleme

120
so call 'PN chirp' technique in 1981, which is the combination of chirp modulation wLth

pseudonoise (PN) phase shift key=rig (PSK/ using a fixed tap matched-filter. A similar

suggestion was also proposed independently by M. Kowatsch et al. [16]. The advanced

version of their proposal in 1983 [16] proposed a PN-chirp spread spectrum system which

combines standard linear FM surface acoustic wave (SAW) devices, DDL, and an external

code modulator. But this proposed system [15] still has the disadvantage of some hardware

complexity and a need for synchromzation, even though it offers high flexibility by avoiding the

use of a programmable matched filter [15]

In the foregoing paragraphs, we have discussed the application of the chirp (or linear

FM modulation) technique in binary digital data communication systems and in various spread

spectrum communication systems. Of tl_e abundant applications of chirp, though, none are

directly applicable for the purposes of our study, because of the complexity in practical

implementation in a satellite link or the nature of the purely theoretical analysis in thOle

proposals. Because of these difficulties, we have investigated some possible techniques using

chirp modulation and PN coding. Ln an attempt to find the most feasible scheme. The desired

features were selective addressing of a large number of potential users operating in a

common band-limited channel, and good system performance measured by low probability

of errors in data transmission

We have reviewed several ways _n which chirp can be used to provide spread spectrum

coding of a communication link. We classified the systems under consideration at large" as

class h 'Within-chirp Modulation' and class-lh 'Multiple Chirp Modulation', as illustrated in Fig.

I and Fig. 2a. The term "Witl_,n Ch,rO Modulation' comes from the fact that a single chirp is

modulated by a finite lengtl_ PN code w,thln a chirp period, whereas the term 'Multiple Chirp

Modulation' comes from the fact that multiple chirps are generated within the finite length of

a PN code.

Also we can further class,N me Wdhin chirp Modulation (class I) as the Amplitude (I-1),

Frequency (I-2). and Phase _-]_ ,e_,o_s tl_at are depicted in Fig. 1. Note that all three systems

need synchronization _n l_e ,_e_er and the anticipated complexzty of this synchronization

2. Review of Chirp SprQ-a S_ Slt_mml

ORIGINAL PAGE IS

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121
ORIGINAL PAGE IS

OF POOR QUALITY

(which was already mentioned in M Kowatsch's proposal [14]) make these systems less

desirable for our purpose of a relatwely s_mple configuration The other catecjory, Multiple

Chirp Modulation', can also be classified as the Multi-chirp (11-1), Time Oelay (II-2), and

Frequency Hop (11-3) as shown _n Fig. 2

The proposed system [I-1 .Mu_tL-ch_rP, is basically similar to the ordinary binary ch_rp

system because it uses up and :o,_vn cn,rps according tO the input digital data. This system

does not need synchronization .n :s receiver Instead. a code correlator which looks for a

specific code word can detect :t_e des,red code asynchronously. For multiple access, the

easiest way is to assign different c_,rp sIo0es to the various users. Another system, 11-2.Time

delay, uses a variable time delay to generate a sequence of chirps w_th time delays set by a

PN code, within each input data 0,t duration. This system needs a complementary PN

sequence in the receiver for sync_romzatlon, and a window with the proper time delay in

order to find the pulses at the decn,q_er Output at the correct times. Therefore synchronization.

and thus the windowing procedure may be the maior difficulttes _n practical implementation

because of their complex=ty ,"he 0mcx_ed system 11-3. Frec _ency Hop. uses the method of

frequency hopping. The transm,ner generates the sequer_-_ of chirps with shifted center

frequencies and this frequency sn,ff n a c_rp stgnal produces a t=me delay at the output of

the dechirper. If the idea of t_ooo,ng ,s apphed for the phase _t _s also possible to design a

similar system to the frequency _>oo,ng case. t_ut this does not apl=ear to be feasible. Unless

the compression filter has exce,_en! _nase ¢ot_erence. comparison of phase at the compressor

output from pulse to pulse *,,,_ _e d,ffiCu=t ;,g2b shows some typical waveforms for the

multiple chirp schemes =llustratl_ _ ¢ 0 2a

Until now, we have 0r_po_ t,, _,nds of spread spectrum CrltrD systems and studied the

possibilities of practical _mO_emenf_l¢_n 'or our purpose in Tdsk 2 If we set the design goal

as a simple structured, sow :_--'.it ear,, _tat_on. systems wh=cn r'eed complex synchromzat=on

can be excluded from co_'_ ce,=_.o_ *'_erefore five of the systems w_th the exception of

Multi-chirp system i1-I ],e opt oe,_ non-starters. Once ,t can be proven that the

'Multi-chirp System' exr,,_'t "e ;,-_o_r :edormance, th_s system seems to be the most

2. Review of Chirp Spre=m $_ Stem*_i _2 2
SPREAD SPECTUM CHIRP SYSTEM ii

class[: within chirp modulation

I-I.AM

I_AII'A ,._

CLOCK

I-2. FM

ORIGINAL PAGE IS

OF POOR QUALITY

eYNC

CI.OCK

DATA

CUT

CLOCK CLOCK

I-3. PM I III I

CLOCK

_ OIFCH *¢1 Pmq

SYNG

CLOCK

OATA

Figure 1. Within Chirp Mo(:lulat,o_s

2. Review of Chirp Spread SOectrum Systems 123
Class_ = multiDle chirD modulation

Tr-1. Multi-chirp

CLOCK

ORIGINAL PAGE iS

OF POOR QUALITY

-2. Time delay

CL()CK

CH I Rill

b

q

cLOCK

"_-3. Frequency hop

dat8 kR l

. ,Jim-, CHIRP

flirt

d

t GIN *

CLOCK

t3GLAY _ A"I'II I/"_1_ d Illll

i

CLOCK

Figure 2a. Multiple ChirD Modulations

2. Review of Chirp Spread Spectrum Systems 1 2 4
CLOCK

COOE'I"

-fl

12-

|I°

Oi

I

I

iii i

t I! I

I

I

\

!

I

i

!

i i

\

I

o o

i !

I

I

I

...--

t

i

i

-L,

I

I I

I i

I i

I

I i

! t

!

!

i

!

I

i

I

I

i

I

/

I

!

I

I

/

I

t

I

I

I

i

i

m I

_meQUeNCy

MOp

Figure 2b. Typical Waveforrni oi _i_i_ml Ci_r!ol.lcxtulati_l

2. Review of Chirp Sprtld $eeco-wm $_rsmml 1 2 S
feasible because it doesn't require additional synchronization in its receiver and it uses the

proven binary chirp modulation scheme which was proposed by Winkler. {7]

This preferred system is depicted in Fig. 2a (11-1), with multiple chirps for signalling. The

signal consists of a sequence of chirp pulses, typically using two slopes, one up and one

down. The sequence is chosen for good cross.correlation and autocorrelation properties, so

that other sequences entering the decoder produce minimal output. Because this system

uses the typical binary chirp system configuration in its modem, the anticipated performance

will be between that of BPSK and FSK, as was pointed out by Berni and Greg [9]. Strictly

speaking, because no synchronization is required, this system is very similar to a

non-coherent FSK system using codewords. However, whereas a FSK system would have a

single frequency narrow band filter to detect a wanted tone, the chirp system uses a "Dechirp'

filter (a pulse compressor) to detect the presence of a wanted chirp.

Let's call this system a 'Coded Multiple Chirp Spread Spectrum System', because the

transmission of a single data bit is achieved by sending a sequence of chirp pulses. The

sequence could be made up of any combination of chirp slo;es, but the use of a single slope

with up and down chirps permits all receivers in a network to use the same pulse compression

filter, with notable cost savings when many receivers are built. Detailed description of the

system is left for the next section for convenience.

2. Review of Chirp Spread Soectrum Sy_ema 1 2 II
3. Coded Multiple Chirp Spread Spectrum System

3.1. Basic System Concept

The block diagram of the proCx_sed system called Coded Multiple Chirp Spread

Spectrum' is shown in Fig. 3a In the transmitting section, the _nput binary data stream with a

relatively slow data rate (compared wllh the chip rate of a PN code) is sorted into logical ones

and zeros by the 1/0 selector at the data input, and encoded with a finite length PN code (18

or 32 chips) with given chip rate Alter encoding, a sequence of up-chirps or down-chirps is

generated by impulsing an up or down chtrp generator (a SAW dispersive delay line (DDL))

These encoded data are summed amplified and transmitted. Because each data bit is now

represented by a sequence of chirp slo0es, the bandwidth of the data has been spread, in this

scheme, a single data bit duration ,s equal to the total number of chip durations of the PN code

word. Typical weveforms and spectra for this system are shown in Fig. 3b. By using multiple

chirps to constitute a code .,or_ t ,s 0osslble to use different slopes for each user. However

the use of single slope w_th uO and oown c_lrp enables the selection of single dechirper filter

(SAW DDL), and thus reduces '._e cost of dev=ce. If all the users in the network use the same

sloped chirp, the multiple acce_t can be accomphshed by assigning different code words

3. Coded Multiple Chirp SI)reul See¢u-,Jm Symem 12 7
OR_GIN_AL P/_.QE HS

OF POOR QUALITY

which promise good autocorrelation and cross-correlation, for example. Kasami or Gold codes

with medium length. [17] [18]

We have modeled the receiving system in two steps: the Dechirper and the Digital Code

Word Correlator. In the dechirper, the received multiple-chirp word entering the receiver

passes through the RF and IF stages and is converted to polar binary data using a pair of

dechirpers and a pair of threshold (Envelope) detectors, tn the dechirping process, the wanted

input signal corresponding to one chip duration is compressed with a predetermined

compression gain and thus has enhanced amplitude. These compressed outputs from a pair

of dechirpers, each assigned to detect a particular up- and down-chirp, are summed (in reality,

one channel output is inverted then summed) to be processed by a sample and hold (S/H)

circuit before code reconginatLon by the digital code word ¢orrelator. Because only one of the

two dechirpers produces a large output at the end of a given chip duration, even in an

interfering situation, the summer g,ves a positive or negative polarized output to the sample

and hold circuit. The sample and hold circuit acts as a threshold detector and passes stable

binary data corresponding to the chip state (one or zero) to _ne word correlator.

The sample-hold circuit output is in the form of polar t_inary pulses, each of one chip

duration, corresponding exactty to the data driving the chirp generators in the transmitter. The

chips have been recovered from the spread spectrum signal without any form of

synchronization, and are always present whenever the receiver picks up a chirp signal with

the correct slope.

The second step is a code word correlator followed by an analog threshold decision

circuit, which determines whether a word is present in the correlator. The dechirper (or chirp

compression filter) and detector behave like a non-coherent FSK system. The digital word

correlator behaves like a un_(]ue word detector used in TDMA systems. Synchronization at the

bit and word level are requtred 'or correct sampling and read-in of the binary word, as in any

other binary data transm_sston system The input binary data from the S/H output and the

self-generated chip sync,_,on,.'at,on clock driven by a Phased Locked Loop (PLL) are

simultaneously given to the :.;.',a[ _ord correlator The digital word correlator consists of a

3. Coded Multiple Chirp Spread Spectrum System 1211
ORIGINAL PAGE IS

OF POOR QUALITY

Coded MultipleChirp.S.S System i

trSn_Itter

< Modulater >

,diUl

< Demodulater >

Figure 3a. Coded Multiple Chno S S. system (proposed)

3. Coded Multiple Chirp $proed Spectrum System 1 2 9
\

i

I

i

I

I

I

l

I

i

i

\

COOS *_*

CO.OOIUlP_ OOT

GO0 "1*

LX:XiiC'Q"

coos

GORRILATOR

OUTDUT PON

OATA i"

Figure 3b. Typical wavefo_t of 3,x:led I,,luttil:le Chirp S.S. System

3. Coded Multiple Chirp $Oreed S_K-.rum Sy_'em 1 3 0
ORIGINAL PAGE US

OF POOR QUAL1TY

pair of word comparators and Modulo-2 shift registers with the same length as the code word.

After the nth (where n is the length of the shift register) sh=ft, the register contents are

compared, and the results are summed and detected by an analog threshold detector, if a

code word is present in the shift register, the output will be ± n. If the spread spectrum is

operated with many simultaneous users in the channel, interference from other users will be

the limiting factor in causing degradation of system performance expressed in bit errors in the

detected data stream. If we use the same chirp slope for all users, the signals must be

separated by the differences between their code words.

3.2. Performance Analysis

The performance analysis Js organized as follows. Section one introduces the system

parameters and waveform definitions for the proposed system. Section two briefly describes

the correlation aspects of up and down chirps; in addition, t,_e cross-correlation of desirable

codes are studied. Gold or Kasaml [17] [18] codes are strong candidates for the code in our

system. Finally, as a general erro¢ cnterta, chip error proba0ility at the dechi_er output is

investigated. Then the probabdity 04 _ (or code) error and corresponding word (code) miss

probability is also described under the constraint of a finite length of code.

3.2.1. System Parameters and Waveform Definitions.

The following are the system 0arameters necessary to define general expressions for

the waveform of a Coded Mu=t,g_e Ch,rD S S. Signal.

Rb; input (binar,/! 3ata _,_ ,ate

Tb; input (binary) data _,=t,ont - I'R=)

Re; Chip rate of the _,cr_ _r :doe sequence)

3. Coded Multiple Chirp Sl=r_et Seectrw_ Slr=n_m

131
ORIGINAL PAGE IS

OF POOR QUALITY

d(t):

G(t);

R,=_,);

R_.b(t);

R,,,,(t);

P;

Chip duration

Chirp slope (or d_s0ers_ve slope)

Processing gain ,'or pulse compression ratLo)

input data signal (b_nary NRZ)

word (or code) sequence with period N = 2" - 1 in: shift register length)

autocorrelat_on of sequence a(k)

Crosscorrelat_on of sequence a(k) and b(k)

Crosscorrelatton of two t_me function

f,(t) and f_(t)

Crosscorrelat_on of code symbol S, andS_

(up and Gown _.n_rc)

Transmitted _ower of s_gnal

Chirp frequenc't sweep range

Then the general express_on of :me wave form for a single chirp duration is given by

So(t) = _'d(t)G(t)cos(o.)_t ° -_-_t 2 _" Co)

where co¢; carrler tad,an frequency

eo; Initial _i_ase t_ raduans

3.2.2. Croucorrelatton o/ Up Chirp and Down Chirp

Because the proposed system uses tl_e chirp techn;que along w0ttl a code sequence to

spread the spectrum, it is necess,=t_ _o _nd the correlation 3etween up and down chirps at

first, even though the relatlon_ f_,O :er_een a series of chirps coded Dy finute length sequences

is needed finally. The autocor,'e0al.o_ jncl ',_e typical spectrum of chnrD are well defined in the

relevant papers by Cock [_91 o,' _, '_ a_oer et al. [1]. So it is ._ot described here.

Assuming only one cn,r_ ,b.C,OQ :uS '_egat=ve for down-chirp. _f _ostt=ve slope is assigned

to up-chirp) is used, the u_-<'_,_ t g,_a_ ,_ g,ven by

U(t) =

e./(oact÷ _ z •

0 2

3. Coded Multiple Chirp $_,_=,=__ Sv=em _3 2
andthe down-chirp signal is given by

O(t) =

eJ(c_ct." "_) I : -_ T _ t _-- T

0 ]tl >7"2

The cross-correlation C(_) between U(t) and D(t) is defined as [20]

C(t) _- < U(t)D(t - :)>

for t > O, and under the assuml_tion the U(t) and O(t) are periodic over T the

cross-correlation C(_) is given by

±_(_ot* "_ e-_:lt _) 2_t- _Jdt ' ,'_ - ' , ,' C(_) ==

= 1T_((%_+ "_I z)_r12, e,[_z2 - _rrldt

- T/2._t

-. 2 flZclt

_+T._÷ t

/'-7 1 ?m

--x) 1Byse.lng.,_, - "2"_ = _, 2

_ej(w=+ ÷ J-W,, _, z

= 4 = 'J?" 2 dX

,/2.r* •

where X, = _'3j:T [1 + Tt

then

1 ._l'J,'," __ 2

C(=) =

44_:T

[,iC(X2) *- C(X_)} + j{S(X2) + S(X_)}]

and

3. Coded Multiple Chirp Sore_ S_ec_rum System 13 3
Ich) l

1

1 [(C(x2) + C(X_)_.2 + (S(X2) + s(x_)_,217

_' 4&FT

where C(X) and S(X) are Fresnel cosine and sine integrals respectively, tt is also possible to

find identical expression to C(_) as ai0ove for the t < 0 case.

For large values of &FT, C(X) and S(X) are approximately 0.5. Then

Ic(=) I = 1 -T_T

therefore, the cross-correlation approaches 0 for large processing gain (or compression gain)

Gr This also implies that the quasi-orthogonality of up and down chirp signals during one chip

duration approaches orthogonality if the processing gain Gp is sufficiently large, Under this

condition, the performance of a chirp system is almost identical to the noncoherent orthogonai

binary FSK system. [2] Fig. 4 shows an example of autocorrelation and cross-correlation of a

chirp signal.

3. Coded Multiple Chirp SgreKI S;)ectrum System 13 4
autocorrelation

-1"

I

/_-F I

,_-71!.r =',.

---_V_ _!r.'!_

itii! -_,,._-w,,I ,_,_, _,

----TOT

CrosscorrelatJon

_A_, i : _,,ll!l,!llA_l!l, Jllilli__ .__ ,, , ,, i ' !, H II I _" _'" i ,' '

T1M_,

Figure 4. Autocorrelation a_<l :,,otP<:or,_auo_ of Chirl_.

3. Coded Multiple Chlr1_ S_ $¢4¢.-ww Sflw. 1 $ S
ORIGINAL PAGE IS

OF POOR QUALITY

3.2.3. Cross-Correlation Between Code (Word) Sequence

For effective Spread Spectrum Multiple Access (SSMA), the following methods can De

considered for the proposed chirp system:

• Using N different codes for N users as in CDMA.

• Using multiple chirp slopes with frequency dispersion variation or time dispersion

variation.

• A mixture of multiple slope and multi-code.

Because the single slope chirp is regarded as the only economical way for simple and

practical implementation, the most desirable method of achieving multiple access is by

varying the code sequences of the network users. Under this condition, the selection of the

codes plays an important role in deciding the performance of the system. The Gold Code [17]

or Kasami Code [18] are strong candidates for our pu_oses, in that they show low

cross-correlation between any pair of codes. Besides, both of them meet the general

requirement of the code sequence that interference due to a d_fferent code be far less than the

code's autocorrelation, and any two codes having the same period must satisfy a lower bound

in cross-correlation [21]. For a detailed description of these codes, see D. Sarwate and B.

Pursley. [22]

For an asynchronous system using code sequences for multiple access, such as the

proposed coded multiple chirp spread spectrum system, the cross-correlation of sequences

is'a key parameter. The aperiodic cross-correlation function of two sequences X and Y is

generally given by [22]

Cx, y,(_) =

N-I -_

_, X_Y/ _. _ 0 < t. < N- 1

,/=0

N-I-_ 1 -N<t._0

E, Xj _ _Y:

j=O

It.I >N

0

where

X, Y; Code sequence hawng same period N

N; Code lenqtn N = 2_ - 1

3. Coded Multiple Chirp Sot•rod Spectrum Syitim I 3 (I
n; Shift register length

f.; Relative time delay between sequences X and Y

and the lower bound of the aperiodic cross-correlation function is

I Cx, y(_.)l < [Cx(0)Cy(0)] _/2

Also, the peak aperiodic cross-correlation magnitude for two sequences is

C c = max[ICx, y(f.)[: 0 _; £ _ N - 1,X =, Y]

whereas the aperoidic autocorretat,on magnitude is written as

Ca == max[ICx_f.)l: 0 < _ _ N - 1l

From the result [23], if there are K users in a multiple access system, with K unique sequences

with the same period N =. 2" - 1, (the Gold or Kasami Codes have period N =, 2• - 1 ),

the following condition must be satisfied to ensure the effective multiple access.

(2N - 1) C2c, 2(N - I) C 2

*

N N(K - I)

> 1

3.2.4. Chip Error Probability (Dechirper BER)

The performance of the proposed coded multiple chirp S S. system can be expressed in

terms of two probability measures c_,p error probability for the dechirper and word (code)

miss probability for the digital code-cos'relator False alarms, in which a data bit is produced

in the absence of a received coaewo_ also contribute to errors =n system performance. In this

section, the probability of err_ ot 'he c_,ps _n the dechirper will be found and discussed.

For the derivation or proolo,=,ry o/c_p error, the follow,ng justifications are necessary.

3. Coded Multiple Chirp Sp_r_ul $1_c:z_m Syfem 1 3 7
.

,

The performance of binary chirp signaling is identical to that of orthogonal non-coherent

binary FSK because the cross-correlation of up and down chirps is negligible, assuming

large compression gain G=

If the justification above is reasonable, multiple chirps, which are derived From a series

of up and down chirps, give a code sequence which can be viewed as a binary FSK sJgnat

waveform

8y assuming the noise is AWGN and that interference from other chirp signals behaves

asa Gaussian Random Variable the generally accepted analysis of non-coherent binary

FSK can be used to find the probability of chip error in the proposed chirp system [2],

[24], [25]

Under the assumption that the above three justifications are admissible, the system

performance, as measured by probability of chip error rate, (denoted by P, or Bit error Rate

(BER)) can be found by either of the following two methods.

(1) Method 1; Direct use of the P. of non-coherent binary

FSK

(2) Method 2: Derivation of the ex0ression for the Po in

binary non-coherent FSK in the presence

of interfering signals

Method 1 gives a simple exoress_on For Po even though it is valid only when the compression

gain is sufficiently large. Method 2 g_ves a more accurate expression without any restrictions.

Here, in this report, only the results are written for convenience. For more detailed

procedures, see Couch [26] for method 1 and Appendix A of Ramanan [2] for method 2.

By method 1, the Po of the dechirper is given by [26]

,D e ----- ._-P

where

p = EJ2N,

E,; average energy per btt

No; two sided noise power s0ectral density

By method 2, P. is given by [2}

3. Coded Multiple Chirp Spread Spectrum System

131
where Q (X, Y); standard Q function

p = EJ2N.

cz = 1/G_

10(X); modified Bessel function of 1st kind,

G_ ;processing gain

Note that the P. of the dechirper derived in both methods above assumes that the signal power

as well as the noise power is the same for each case of up chirp and down chirp. In addition,

noise is assumed to be AWGN and interference used in method 2 is assumed to be a Gaussian

Random Variable.

Fig.Sa shows the P. of the dechirper when it is calculated by method 1 assuming AWGN

in the channel, and that the dechirper is identical to a non-coherent binary FSK system. Fig.Sb

[2] shows a comparison of the calculated P, curves by method 1 and method 2 for several

compression ratios. From Fig. 5b, it can be seen that for large Time-Bandwidth products,

hence for large compression ratios, non-coherent FSK (orthogonal) and chirp signals

(quasi-orthogonal) have nearly identical performance.

3. Coded Multiple Chirp Spread Spe_rum System 13S
ORIGINAL PAGE IS

OF POOR QUALITY

p -* _ •

• 2 I S_

C_qlP _ _ ! al' Ol[l_llll_.(Nlal 1'0 teal' ¢811E.._I

Figure 5a. BER [Chip Error P-_ol _.f 3e<:_,r_u'. [Prior to Code Corretatorl

3. Coded Multiple Chirp 5pro,.l Spe_rum System 1 4 0
CHII=kP MOOULATION BT=400

¢

uJ

>

_H

I0 "s

NONCOHERENT

FSK

\

\

\

\

\

\

\

\

\

\

\

CHIRP MOOULA'_ON ST= IO

Figure 5b. Probability of _r_ ,w_s ED, N0 for noncoherimt FSK and chirp _gnall of different

time-bandwidth products.

3. Coded Multiple Chirp Spiel SlNK_rr_ |TIII_ 141
3.2.5.

ORIGINAL PAGE IS

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Performance of Digital Code Correlator

A digital code correlator ,s used to determine whether the received code (word)

sequence is matched with the un_aue word stored in the comparator. Because the proposed

digital code correlator uses the same dea as a unique word (UW) detector used in TDMA

systems [27], the performance of our corretator can be directly formulated from that of a UW

detector (correlator) used ,n '._e ,_TELSAT system {28]. Because two identical code

correlators are used to detect a _,nar'y 0' and binary '1', the probability of a word (code) error

for an individual code (word) corre_ator ,s 'ound initially, and then the joint probability of error

can be determined.

Let's find the probab_l_t'y ol ..,on:l error Po for a single code correlator by the result of

Feher [29]. The probability tha! an N 0,! !crop) word is received correctly is given by [29]

P :" i_=o(_1 pt(1-p)N-'

where N is the word length. E ,s the _um0er of errors to reac - me code correlator threshold.

'p' is the bit (chip) error pro0ao.,w .._1c_ was previously defined as the P. of dechirper in

section 3.2.4. In addition "r ,s T_e _u_oer of 0ossible error b,ts recetved from the dechirper,

under the constraint that I < E

Because the probabd=_ ot co_'ect ..otd ctetectlon =s gtven by P the probability of word

miss Q, for a given threshold E s _;_e. _y

Q = 1 - P=l--_-' ' :'""

The above proba0_l,ry _ _ "-.Is Q ,s a function of code _ength N and dechirper b_t

(chip) error rate, so there ca_ :_ _,e,,_ :_ :ases to calculate. In _,g 6. the 0robabdity of word

miss Q as a function of dec".,_, i:J,il 0 s ,_ustrated for N = '6. 32. 64. and 128. with E =

3. Coded Multiple Chirp $ar_e_ Ile,_9-_= S_m _4 2
ORIGINAL PAGE IS

OF POOR QUALITY

O, 1, 2, and 3. It is seen that code miss rates can be made quite low, even for fairly h=gh

bit-error rates of the dechirper, especially for large value of N [29] However medium length

codes like N = 16 or 32 are recommended because of shift register size restriction.

Another measure of system performance is word false alarm rate F A word false alarm

occurs when bit patterns due to random noise are allowed to enter the code corretator. There

exists a small but finite probability that random patterns will accidentally correspond to the

unique code and cause the erroneous production of a correlation detection. The probability

of a false alarm F is given by the probability of accidental occurrence of the unique code in the

wrong location within the number of errors E established by the unique code threshold and

expressed by [27]

Fig. 7 illustrates the dependence of F on code length and _t (chip) error probability of the

dechirper. In this figure, umcZue code false alarm proba0_lity, F, and unique code miss

probability, Q, are drawn as a function of selected bit (chip) error rate and code matching ratio

M/N, where M is the number of b_ts _n the received sequence of a code that must match the

unique code stored in the correlator [27] (M is the threst_old of the word presence detector

which follows the correlator and _s mggered by the correlation between an incoming bit

pattern and the stored code sequence )

3. Coded Multiple Chirp $pr=_ $oectrurn System I 4 3
b-_ E-,' ;_- , £ -4

Figure 6. Word Miss Probabd_ty vs Oechirp_" BER.

3. Coded Multiple Chirp Spread Spectrum System 14 4
E--1

WOPlO _ALSE ALARM ,=NO _4tSS DROSAIIILII"ES

ORIGINAL PAGE IS

OF POOR QUALITY

Figure 7. Word False Alarm P,'oOao_J,ry vs k41SS Probability

3. Coded Multiple Chirp Si)rq*'* $1_rr_m Symim 1 4 S
3.2.6. Chirp Spread Spectrum Multiple Access

As was briefly mentioned earlier in section 3.2.3, the method of varying the chirp slope

either by time dispersion or by frequency dispersion, or using a unique pseudo-random code

can be employed for multiple access with chirp spread spectrum. In addition, hybrid versions

of those methods can be considered. In other spread spectrum methods like Direct Sequence

(DH) and Frequency Hopping (FH), the number of users who can access the transponder

simultaneously is limited by the cross-correlation achievable between the codes used. tf the

method of slope variation is used, the major decision factor for the number of simultaneous

users (n) will be the cross-correlations of different sloped chirp waveforms.

The signal selection to keep low cross-correlation between the multiple simultaneously

chirps must involve taking signals with differing time-bandwidth products, because the actual

difference between the products determines the degree of orthogonality. Overall system

performance depends on the influence of all the interfering s;gnals on any one channel. From

this point of view, the number of simultaneous users is dictated by the following three factors,

regardless of slope varying method:

1. The time-bandwidth product of individual signals

2. The minimum difference =n the time-bandwidth products Used

3. The summation of n-1 distorted outputs has to be below a certain desired level for any

given user for satisfactory o(0erat=on.

The analysis of cross-correlation {2] between a desired chirp slope and an unwanted

slope confirms the dependence of cross-correlation on the above mentioned time bandwidth

product. Therefore, proper adlustments between slopes may allow multiple access to a certain

degree. However, this schemeLJs_ng d_fferently sloped chirps is not considered a good choice

3. Coded Multiple Chirp SprllKI Spectrum System 146
for practical applications, because varying time-bandwidth products result in a different data

rate for multiple users if the time dispersion is varied, and different bandwidth utilization in the

limited transponder bandwidth. In addition, different SAW devices are needed for all the

multiple users. From this point of view, spread spectrum multiple access using chirp with

varying chirp slopes seems to be a bad choice. This conclusion is in agreement w*th

reference [2], where it was shown that only 5 users could share a typical transponder using

multiple slope chirp.

If we use the method of assigning unique codes for multiple users, while keeping the

same chirp slope for all users, mult=ple access is possible. The performance of this scheme

is no different from other conventional spread spectrum techn*ques, and the merit of a

relatively simple implementation in the chirp system will be retained. The alternative choice

of combining slope variation and a unique code also seems undesirable because the system

gets complicated. In conclusion, spread spectrum multiple access using many slopes in a

chirp system is not considered desarable, even though it is possible, compared with other

spread spectrum techniques. In this s=tuation, spread spectrum overlay on an existing service

seems to be the proper alternattve to keep the inherent advantages of chirp.

3. Coded Multiple Chirp Sate "_ $_N:=rtm_ Sye=em 14 7
ORIG_qAL PAGE IS

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4. Overlay Service -An Overview

Spread spectrum overlay is a ktnd of practical implementation of "frequency reuse" by adding

a comparatively low data rate signal to an ex=sting high traffic channel. This service can increase the

capacity of an existing satellite transponcler, hence can provide additional service and function to

existing users in the satellite network Because the primary concern is the effective reuse of the

existing system without causing any harmful interference to the users already occupying the

transponder, minimizing the possible occurrence of mutual interference between the existing signal

and the overlay signal is the key factor tn th=s service.

The performance degradation of existing signals caused by overlay can be kept to a minimum

by reducing the transmitted power of the overlay signal. The data rate of overlay signals then must

also be reduced to maintain the desired bit energy to noise density ratio. In this situation it is obvious

that the overlay signal experiences re(at_vely high level interference from the existing signal, and

spread spectrum techniques are a :ro_er choice tO reject this interference, thereby achieving

effective frequency reuse or overtay

In the study of this task. s_read s_ectrum overlay, the existing signal being sent through the

transponder is confined to analog r._,_-_ Other possible choices such as FDM telephone and data

signal were not studied. FM-TV _s _rc0a01y _he most susceptible signal for interference from a chirp

4. Overlay Service -An Overview 14 8
overlay, because both signals are frequency modulated, so the FM-TV demodulator is capable of

demodulating the chirp signal directly.

This report is organized as follows. Chapter 5 introduces the fundamental analysis of

interference between analog FM.TV signals and chirp signals by modeling the FM-TV signal in various

forms. Chapter 6 describes the procedures of modeling the FM-TV signal in practical form by using

multi.level step functions, or random multi-level step functions. In these procedures, some resuits of

computer simulations are illustrated and analyzed both in the time and frequency domains. Then,

based on our interference analysis between television and chirp s.s. signals, a satellite link analysis

for C band is performed to find the proper C/I ratio in a practical situation. A comparative study of C/I

ratio and optimum overlay position within the analog FM-TV bandwidth, and a performance analysis

of an analog FM.TV system with chirp overlay concludes this report.

4. Ovedty Service --An Overview 14 9
ORIGINALPAGEIS

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5. Interference Analysis I (General Approach)

For effective overlay service between existing analog FM television and spread spectrum chirp

systems, mutual interference between the two s_gnals must be studied. It is first necessary to define

the type of signals used and the recetvers for these two systems. What is needed is a good

description, in the frequency doma=n, or a FM-TV signal. Unfo_unately, there exists no finite

description or definition of this form that we could rind in the literature. Therefore, modeling the TV

signal appears to be a major difficulty tn our study.

In this chapter, the demodulation 0rocess in analog FM television (TV) and the proposed

spread spectrum chirp system are briefly reviewed as a preliminary stage. Then, for a defined chirp

signal and frequency discriminator _n tree 1"V recmver, various case studies of interference into the

analog FM-TV channel were conaucted Js,ng an analytical approach. As another phase of the

interference study, the interference :aused _y FM-I"V signals into the chirp channel was also

investigated. In this analysis, because *e ',ad no definition of the FM-TV signal and its corresponding

spectra, computer simulations as well is 3nalyt[C expressions were used to support the analysis.

5. Intarfaranca Analysis I (Ganatig Aoo¢oe(_! 150
5.1. Analog FM-TV demodulation

The process of FM-TV demodulation recovers the baseband signal as a TV video signal. A

frequency discriminator or PLL demodulator, which follows the receiver IF stage, is used for this

purpose. Only the frequency discriminator _s considered for our purpose. The received FM-TV signal

is converted into baseband form after RF amplification and IF amplification. The FM discriminator is

usually followed by a deemphasis filter and a video amplifier. In the situation when there is

interference added to the FM.TV signal, an error voltage will appear at the discriminator output

according to the intensity and phase relation of the interference signal to the FM-TV signal. This

phenomena will be explicitly dealt w=th _n t_e next section.

While interference to the FM-I"V signal reception can be analyzed through the

FM.discrtminator output, the other aspect of _nterference in an overlay service is analog FM..TV

interference into the chirp signal c_annet of the proposed coded multiple chirp spread spectrum

system. Because the interference power from the FM-TV signal is generally at a much higher level

than the received chirp system, the performance of the chirp coml0ression filter and the receiver IF

filter preceding the dechirper input pray an _mportant role in this interference analysis. Obviously, the

output of the dechirper filter (compressor) must spread the input interference signal into a wider

duration of time for effective detection of tl_e compressed chirp pulse.

5.2. Chirp interference to analog FM-TV signals

In section 5.1, the principle or ,be _e_<lulation process of an analog FM-TV signal was briefly

reviewed. Because the FM-discpr,',"=tor 3utDut =s expressed as a voltage, the response of the

discriminator of an FM-TV recel_e_ '_, ,ar,oul ,nputs must be studied before practical modeling of

S. Interference Analylil I (Generta AlH),_e¢_) 1 S1
analog FM-TV signal can begin, In this orelimlnary analysis, we found a generalized formula for

FM-discriminator output with a general message s_gnal which frequency modulates tne car,Ter Th_s

formula can be applied to any kind of message information, video or audio, that is used ,n an ._M

system. Fig. 8 and Fig. 9 illustrate examl31es of the time waveform of F:M-discriminator output error

voltage for an unmodulated 17/carrier and a modulated carrier with chirp interference, under various

input signal to interference ratios.

5. Interflrlncl Anafl/ei8 1 (Gene_ll Aim,-,m_) I S2
=:

:3.

II

m_ m

Ibo

':7 v

°3<o

"rl &3o0

(-

13

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¢:3g

¢-

11

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egb

(3

3"

-2.20

0

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PEAK VTG [V] I E+L

-0. io o.oo n.io 2.20
88

mIAE6EN

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Lit_ I

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< 1 > General Expression of FM-discriminator Output Under Interference.

To see the effect of interference to the FM signal, we need to find general expressions for error

phase, instantaneous frequency deviat=on and corresponding error voltage at the FM.discriminator

output caused by the interference.

If the desired FM signal is expressed as:

w(t) '= Acos [colt ÷ q)(t)] (I)

where A: FM carrier amplitude

co_: FM carr_er radlan frequency

q)(t): FM career phase

and interference is expressed as

A

i(t) '- _ cos[o_t 4- _v(t) 4- _]

where R: interference canner amplitude

co=: interference carrier radian frequency

_v(t): interference carwer phase

A

I_ ' initial phase

then, the input signal to the FM-discr_m,natot ,s the sum of the above two signals, written as

S(t) = ..,,tl - ,{t}

- A,T1cosi_It + _(t) + I](t)]

where AZ(t) = (A - _ :_{'D_tll}' " _R stn[_(t)]) '=

(2)

(3)

lss

S. Interference Analysis I (Generad Aam*_e¢_l
^

• (t) = ¢odt + _v(t) - q_(t) _- _.

cOd = (02 -- (o1. carrier frequency difference

_(t) ' error phase

In detail, the Error phase caused by interference will be

A

_(t) _ tan _ 1 I R sin[(_°dt "_ _J_(t) i (_(t) JP J_] 1,_ ,,,= A - Rcos[codt + _v(t) - q)(t) + I_]

(4)

and the corresponding instantaneous frequency deviation is

d _(t)==(-_-) _'(t)[_- + cos (:l)(t)] (5)

• _f =' dt E1 + (_.)2 . 2(AR.) cos_(t)]

therefore, the error output voltage for the FM-discriminator with slope kD is

<2> Complete formula of FM.discr_m_nator Error Outt0ut

Based on the general expression ,n Eq. (6/ above, the general case FM-discriminator output

error voltage for any FM-TV career and energy dispersal signal and linear chirp interference can be

written as:

5. Interference Analysis I (General Al=_ro=c_) 1se
So(t) =

• , ACH +cos(._l_t 2 + cl2(t)

I (ACH) [_..... d : __t22.._'_dm(t_ ) +_d._2(t_ ) - dIZt)![AT.--_V.-_ .....

kc) ATV rl + ( ACH)2 + 2(ACH) cos(1,t 2 _- d2(t) - 2_fd j-t m(c_)da -

L AI"V ATV z

_. 2t_fd j't m(¢)dn - dl(t)

dl(t) + _dl+.p.

where

ATV ; W-carrier amplitude (cf. W(t) = ATV cosEo_It + q)(t)]

ACH ; chirp interference amplitude (cf. _(t) = ACH cosEo_2t + -_t 2] )

o)1 ; TV carrier radian frequency

°)2 ; chirp carrier radlan frequency

(Od = ¢_2 - o)1 ; carrier frequency d_rference (radian)

K D ; FM-discdminator slope [VolPHz]

p; chirp slope [Hz/sec]

fd ; TV signal peak frequency deviation [Hzl

re(t); active "IV video signal (message s,gnal)

dl(t ) ; Energy dispersal signal used in "rv stgnal

d2(t) ; Energy dispersal signal used _n ,nterference (chirp)

q)(t) " 2_fd _t m(ct)do.; "IV carrier phase

• initial phase offset of interference.

NOTE: This expression is the same as me So(t) of case 4 described in detail in a later page. except

for the existence of initial onase ;= ,n cmro _nterference term.

<3> Case Study of Interference E_ec_

S. Interference Analysis I (General A_),_4¢_) 1S7
Withthe formula previously derived, the effect of chirl_ interference on a TV signal in various

cases was studied as follow:

1. Case 1. [No interference, used as a reference case]

When there exists no interference, there will be no phase error, thus no frequency deviation and

no error output voltage from the AM-discriminator.

input to FM-disc; Xr(t) = ATV cos[e_lt + q_(t)]

where q_(t) = 2=fd Jlm(cc)d¢

phase error ; 0

instantaneous frequency deviation: 0

FM-disc output error voltage • 0

2. Case 2. [unmodulated carder plus ch_rD interference]

a. input to FM-DISC;

1 2

Xr(t ) = ATvCOSCOlt + AcHcOS{CO2t + -_t ]

b. phase error;,

_(t) = tan

1 t2, 9

-I[ A AcHs,n,_,..jt "- _+ I) 2// TV _ ACH cos lobar -_-_t )j

c. instantaneous frequency de_ at_on

S. Interfaran¢l Analysis I (Genartl A;)proa¢l_) 15 II
Af =

AcH(°_d + IJ't)IA*C_H "+'L'_TV cos (--_-p.tl2 + _ctt)l

ATV1 +" _ + '_TV ('_'Lt2 + c°dt)

d. FM-disc error output voltage;

I KD (ACH )

NOTE: 1. As a first approximation, assuming ACH < < A.rv then,

_ 1 KD ( ACH ) cos(-_-pt 2 + o)dt) So(t)app r°x 2at ATV (°_d + I_t)

3. Case 3. [Frequency Modulated "IV Carrie_" plus chirp interference}

a. input signal to FM-DISC:

'Xr(t ) = ATV cos{o_lt + 2aftd St m(a)da] + ACH cos [o_2t + -_-I_t2]

where 2=fd jl m(a.)da = (p(t)

re(t): active I"V message signal

b, phase error;,

= tan-iF ACH sin [y(t)]

LATV ÷ ACH cos [y(t)]

where y(t) == _/CH -_;_TV _" COclt

5. Interference Analysis I (General Approach) 1 SS
_ 1

2 #t2 - 2_fd !l m(_lda + COdt

1 2

_VCH = --_-p.t ;chirp ,nterference phase

C. instantaneous frequency deviation:

[ed + p.t -2zf_m(t)i _ ÷ cos{y(t)]

1 + (-_--, AC_ ,' - 2(A.1. _C..__..__H V ) cos{y(t)]]

d. FM-Oisc output error voltage:

1 KD ( ACH ) .

S°(t) = 2--'_" ATV [cod _t - 2_fdm(t)].rl L

ACH + cos{y(t)]]

ATV

ACH = ( ACH ) cos[y(t)] ]

+ (-_--_--) + 2 AI.V J

Note: For approximation of So(t) as=umsf_g ACH < <ATv, use !be following formula:

8o(t)approx = _K O (_----_-)[oo d " _ - 2*fdm(t)l cos{y(t)l

where 1 2

¥(t) = -_t - 2_feJ'Tm(mda + (odt.

4. Case 4. [Case 3 with an added e.e,_M _,toe,'_al s=gnat]

a. input signal to FM-C)ISC:

Xr(t ) = ATvcos col t o ;,_,j, "-,=_o= 4- dl(t ) 4- AcHcos co,2l _- -_-p.t

b. Phase Error;,

5. Interference Analysis I (General _ 1e0
_(t) = tan-1 f ACHsin[Z(t)] t ATV + cos [Z(t)]

where Z(t) = q)CH d - q_TVd

=-_d.Lt 2 + d2(t) - 2,_fd]'tm(a)de,- dl(t ) + codt

_FCHd = _VCH _ d2(t)

q)TVd = (p(t) + dl(t)

dl(t), d2(t ) • Energy dis0er_al signals

c. instantaneous frequency deviation;

&f == ( ACH )

ATV [,.Ac. )2 c.,z(,,,]

where _ d dl(t ) d_(t) - d d2(t )

d'l(t) dt d-'t"

d, FM-DISC output error voltage:

1 _, , ACH

So(t) = 2"-£_'°_'_'Ew)

' " l[ ACH ÷ cos [Z(t)]] [_d ° _ - 2=fore(t) + d2(t) - d_(t)JL A'rv

[1+( ACHA1.V)2+ 2( _-_--_-) cos [Z(t)]]

NOTE: For approximation of So_tl =ssum,ng ACH < < ATV, use the following formula:

I I

I KC) (ACH , ,_ ,- _t - 2_fdm(t) + d2(t) - d1(t)lcos[z(t)]

S°lt) = 2"-'_" -_"-_'-

$. Interference Analysis I (GenertJ Apl_,t_) 1 81
where

1 2

z(t) = _-#t

ORIGINAL PAGE IS

OF POOR QUALITY

t

4- d2(t) - 2rtf d f m(_)dcc - dl(t) + (_d t

5.3. Simulation of FM.discriminator output; chirp interference

case

In the previous section, some useful basic formula were found for the FM-discdminator output

in the presence of chirp interference tn addmen, _llustrations Fig. 10 _nd Fig. 11 show the simple case

of the error voltage output of an FM-d_sc_.m_n'ator when the FM.TV ._-essage signal has a simple form

like an average d¢ voltage. In this section a more complicated but _ore general form of the FM-TV

message signal is used as the T'V message signal, to determine a realistic result for the

FM-discriminator output. Even though a more practical form of TV message signal with preemphasis

filtering is simulated in a later part of tl_s rel:)Or't, this section presents results for an interference

analysis with a simpler form of signal

In the simulation here, the reference 1"V _tdeo signal which we are using is a staircase voltage

between 0.25 and 0.75 volts, with durat,on of ]5 _,S On a TV screen th_s v_deo signal would produce

a series of vertical bars of decreas=_j 3r_gr_tness wcth a seven step gray scale. To test the accuracy

of the simulation, a staircase s,gna= ,,as :o_verled into the corresponding FM-wave. without any

bandpass filtering, and then processe_ "_'ougn t_e receiver simulation The resulting video output

was plotted, giving the result in ;g '0 _ ¢.g _1 Fig. 10 shows the transmitted staircase function

and Fig. 11 illustrates the video :_:,,t _e_ ow pass filtering w_th 4 2 MHZ cutoff frequency, the

theoretical bandwidth for NTSC • _eo _ ;,,]_s _ote ',hat ,n Fig _1 lhe transients caused by the

5. Interferencl Analysis I (GeneraJ Aee_¢_l _S 2
ORIGINALPAGEIS

OF POORQUALITY

instantaneous step in the original signal in Fig. 10 cause the receiver filter to ring at the expected

frequency. In the above simulation, the FFT routine used 2048 samples of the 64_s waveform This

sampling rate will theoretically reproduce all FM signal components up to 16 MHz.

The chirp signal was then overlaid on the FM-TV signal, and the error in the video staircase

waveform at the receiver (FM-discriminator) was calculated and plotted. As mentioned earlier, when

sampled at 2048 or 4096 samples in 64 _s. alias=ng does not appear and the resulting interference _s

small. Fig. 12 shows the error waveform for a 26 p.s chirp signal swept through 10 MHz. The chirp

signal was set 26dB below the video s_gnal, measured as a peak voltage ratio. The largest

interference occurs at the end of the chirp signal, with a peak level of 0.005V, relative to a 1.0V nominal

video signal. It is unlikely that this low level interference would be visible on a "IV screen, and would

appear as very small random dots if rt could be seen. To check whether the chirp signal produced

significant interference in the video s=gnat spectrum between 3 and 4 M-z, where the chmminance

information is transmitted, additional s=mulations were performed. ; 13 shows the resulting

spectrum of this interference component For comparison purposes, F, 14 shows the calculated

spectrum of the video staircase signal. The interference power levei is well below that of the video

signal, and has a very fiat spectrum, _nd_cat;ng that chrominance inte_erence should be very small.

From the above results, there appears to be very little interference at the video output of the

TV system when a chirp signal is overla;d on an FM-TV signal, as carried by many satellite

transponders. The interference peaks are more than 40dB below the FM-TV video signal when the

chirp RF signal is 20dB below the FM-I"V career, showing good interference suppression by the FM

carrier. In the simulation procedure, the effect of deemphasis in the video receiver was ignored for

convenience at this time. In general, deempnas=s in the receiver will help to reduce the effect of

interference peaks because of its low pass 51ter action. However, in a later part of this report, when

a practical interference model is used _reemphasis in the transmitter was added to get more

practical simulation results.

5. Interference Anllysil I (General AOgro=¢_l

163
I I I I I I

I-.

_1

0

w

n

s8

L_

L_

2.42 3.21

TIME(SEC)

4.01 4.81

E+S

G. 40

Figure 10. Simulated 9 step staircase function (time plot)

5. Interference Analysis I (General Approach) 16 4
[N=2048.FC=4.2_IHZ.UNIT GAIN]

o

=.==-_

I I I I

0.80 1.60 2.40 3.20

TIME [SEC ]

4.00 4.80

E+S

I

i

S.SO 6 40

Figure 11. Received and Iowpass filtered sDmulated 9 step staircase function

5. Interference Analysis I (General Approach) 1 (IS
T T T

$

I

8

÷ I

0.00 0,80 | .60 2.40 3.20 4.00 4.80 6.60

TIME[SEC] m E+S

6.40

Figure 12. Error output voltage of FM-d,scr_mmator after bandpass filtering

5. Interference Analysis I (General Apgroach)

166
O

!

!

iI,i

_

!

I I I I I

I

0 13 26

I I

[BW=I.0M

61 64

FREQ [o

4 I

?9

12

9O

1G [ MHz ]

Figure 13. Spectrurn of error output after I_mClOa_ur_ filtering (3-,4 MHz)

S. Interference Analysis I (General ApI_4(/_I 1(i 7
o

r,,,

m

!

_llJ_

I

@

!

0

| I I I I I I

m

1

13 26

4 8 12

FREQ [O • E-I

102

16 [MHz]

Figure 14, Spectrum of 9 step staircase functKon

5. Interference Analysis I (General Approach) 1 6@
5.4. Analog FM-TV signal interference on chirp demodulation

As another phase of the mutual interference study, the analog FM-TV signal can be modeled as

interference in a chirp signal overlay service. To see the effect of such interference, we evaluated the

output of a dechirper (compression filter) for the desired signal (chirp) and for the interference (FM-TV

signal). For simplicity, detailed procedures are omitted, and only the results are described here.

Let the desired chirp signal be:

f(t) =

ej(coct+ 2Pt 2) -T o < t < To

0 Itl >T o

and the unwanted interference analog

i(t) =

FM-TV signal be:

AieJ[_°ct+_(t)] I o

-T 1 <t<T 1

elsewhere

then the compressed output of the desired chirp signal is given by

/2_T2 sin/_tTo .. t 1 t2 _r

eJL(Oc - _-p. + _']

g(t) = v/ _,_ _tTo

and its real part is written as:

/2_T 2 sin/_tT o 1 2 _r

Re[g(t)] = _ #tTo cos E_ct - -_-]

5. Interference Anelyuia I (General Approlch) 169
whereas the compressed output for analog FM-TV interference is

/z ej(_oc t + _r T1 dteJ[(_(T) ___.(t_T)2]

J(t) -- AiJ_ _') /" T1

and its real part is written as

Re[J(t)] : tA i 2W/2_ .f_TT1 1 cos [(_c t + -_-) -- _-('r--t) 2 + _(_)]d_

In all the above equations, #, coc, _(t) are the same as in the definitions in section 5.2. Based on

the equation for the desired output g(t) for chirp and unwanted outputs J(t) for analog FM-TV

interference, a computer simulation was performed under the following conditions:

chirp frequency dispersion

chirp time duration

chirp dispersion slope

FM-TV message signal

FM-TV signal duration

10 0MHZ

26 _,s

10 0/260 MHZ//_S

0 SV dc

64 ,,S

FM-TV signal.max frequency dewahon 10 0MHz

Initial IF frequency 70 0 MHZ

interference signal amplitude 0 _V 02V. 05V.

For convenience of simulation, the message s_gnal of the analog FM-TV was chosen as 0,SV dc,

which is the average dc level of a v_de_ s,gnal =n a conventional analog FM-TV system. Fig. 15 and

Fig. 16 show the outputs of the compress,on filter Fig. 1S is the typical output of a compressor to the

desired chirp signal. Fig. 16 illustrates 'me output of the compressor when s_mulated FM-I'V signal

interference with an amplitude 0 ;v Jc 5 apphed. As can be expected from chirp theory, the

mismatched (unwanted interference_ sgnal .s w_dely spread into a long time duration, while the

matched signal (chirp) is compressed _o • snortttme. From this result, Jt is possible to expect that

5. Intorfarence Analysis I (General ,&oOtee¢_l 1 7 0
the interference from the "FV signal can be effectively suppressed below a certain desired level if the

proper conditions are )rovided in the overlay service.

S. Interference Analysis I (Gener_ &_M_I_) 1 71
8

! I I ! !

[INPUT=CHIRP, DU=26US #=0.3846]

Figure 15. Typical compression filter o_T_t _, cn,q_ s_gnat input

6, Interference Analysis I (General A_I 1 7 2
8

$

0

! i

1 I

Figure 16. Compression filter output _o¢ cM._# srgnal with amplitude of 0.1V

5. Interference Analysis I (Generat A;)oro_) 1 7 3
6. Interference Analysis II (Practical Modeling and

Simulation)

So far, we have investigated the principle of an overlay service between analog FM-TV and a

chirp spread spectrum system by finding the general expression of the FM-TV demodulator output

error voltage when a chirp signal is interfering with the FM-TV signal, and vice versa. In those

procedures, the message or information signal of the FM-TV has been simply modeled by adc voltage

or a pure sinusoidal wave, to investigate the general case of an interfering environment. However,

it is essential to define a complete model of a "IV video signal in the time and spectral domains, for

the further analysis of an overlay service.

In this chapter, the modeling procedures of an analog FM-TV video signal without

preemphasis, and corresponding computer simulation results are explained first. Then, the same

procedures are repeated with a preemphasis filter. Finally, an overlay service link analysis for C band

is clone to permit a performance analysis in section 5.

Precise modeling of a general TV video signal is almost impossible because the spectrum of

a general television signal is not known with any degree of certainty The variety of images being

shown gives a variety of spectral shapes tn addition, the FM television signal is a wideband signal;

in practice the entire overlay signal (chtrp) will be contained within the bandwidth of a television

6. Interference Analysis II (Practical Modeling and Simulation) 1 74
receiver [31]. Nevertheless, a realistic model of a television video signal is required for analysis of

the case of a _ signal overlaid with a chirp signal.

As a model for computer simulation, a typical television video signal with NTSC 525 line format

was selected. The usual television signal contains synchronization pulses (vertical and horizontal),

audio, color bars (in color television only) or chrominance patterns, as well as the luminance signals.

In horizontal line scanning, about 64 _.s duration (63.55 _.s exactly) of a single horizontal scan duration

contains all the necessary information. Of the 64 p.s duration, the first 11 p.s is devoted to

synchronization and the last 13 p.s is used for color burst in a color television. Therefore, the active

video signal uses only 30 ~ 40 p.s duration in a single horizontal scanning period of 64 p.s. The

maximum video signal is 1,0 volt peak to peak amplitude.

6.1. Analog FM-television signal modeling I- Without

preemphasis

A. Linear Step Staircase Function Model

Based on the typical model described above, a simple model in the time domain is illustrated

in Fig. 10. The corresponding time dommn exloression can be written as a series of staircase functions

as follows:

m(t) = AIU (-t+_l) +" A2U(-t+_2) + ....A7U (-t+_7) +Asu(-t+'r8)

or, in general form, it also can be wrItten

6. Interference Analysis II (Practical Modeling and Simulation) 1 7 S
rl

re(t) = _'AiU (_t_i):n=l .... m

i=1

where m(t); analog FM-TV video stgnal in a single horizontal scanning period (64 #s).

As; incremental magnitude of unit step function in voltage, 0 < A i < 1

"ri;time increment of luminance signal corresponding to magnitude A i

m; desired number of luminance level.

As an extension of this model, the step size was increased to 200 later, as in Fig. 17.

6. Interference Analysis II (Practical Modlllin 9 and Simulation)

176
VMAX=I.O.TMAX=S4US

I

un0lased

0

°0.00 0.80 ! .SO Z.40 3.20 4.00 4.80

TIMECSEC] _ E+S

S.SO

Figure 17. Time plot of simulated 200 steO sta=rr.ame function model for video signal

6. Interference Analysis II (Practical Modeik_ =_1 Simulation) 177
B. Random Step Staircase Function Model

The model of an analog television video signal with a linear, 9 step or 200 step staircase

function can be used to represent a typical video signal, even though it is simple. But this cannot be

the worst case model which can be encountered in a real system. So another model was made which

consisted of 256 steps of random amplitude. A uniform random number was generated with an

arbitrary seed, in the range of 1 to 256. Incremental times between 17 #s to 50 #s were given random

amplitudes which were generated by the random number generator. This gave a random amplitude

step function. Because of the uniform distribution of the random step amplitude, the mean value was

0.5 Volt, as with the linear staircase function Fig.18 shows this function.

6.2 Simulation of Wideband FM-TV Signal and Chirp I-No

Preemphasis

As mentioned earlier, an analog FM television signal is a wideband signal. In domestic

satellite television transmission, the basedband television signal, with its video information bandwidth

of approximately 4.2 MHz, is FM modulated so that its RF bandwidth fills an entire 36 MHz bandwidth

of a satellite transponder [2].

In the overlay service, when a chirp signal is overlaid on an existing television signal, the

interference of the chirp signal into the telev=s_on signal is an all pass function because the chirp RF

bandwidth is usually much narrower than that of the FM television signal. But in the case when the

chirp signal is interfered with by the _,elev_s_on signal, only a specific 50 MHz bandwidth of television

interference power is introduced in the chirp s_gnal via the tF filter (if its bandwidth is set to 5.0 MHz).

For our study of mutual _nter_erence m an overlay system under the above conditions, the

spectrum shape of the televiston sgnal =s modeled by choosing a staircase or random video

6. Interference Analysis II (Practical Modelln 9 and Simulation) 17 $
waveform.Computersimulationisaneffectivealternativeto thepracticalexperiments,eventhough

the finaljudgmenton mutualinterferenceis highlysubjectiveanddependson the response of the

human eye in the television case. Because of the complexity of modeling RF signals, all the

simulations were clone in baseband. By trial and error, we found that 2048 or more samples in the

64 p.s horizontal scanning period guaranteed no aliasing effects. So 2048 samples were used for

overall simulation, both in the time and frequency domains. The chirp signal used in the simulation

was the positive (up) frequency swept signal with 10.0 MHz/26.0 p.s sweep rate. By using 2048

samples, the maximum frequency span was -16 MHz to +16 MHz.

In spectral illustrations, the frequency i.d (or index) is the multiplier of incremental frequency

(when N=2048, Af = 0.015625 MHz), so, for example, the index number 100 means 1.5625 MHz. The

the vertical axis of the graphs is the power density of the signal calculated with its minimum level set

to -100dBW or -120dBW, assuming the maximum level of carrier amplitude is ,_" for future

convenience in power calculations.

For baseband simulation, the transmitted analog FM television signal and chirp signal are

defined as follows:

t

w(t) = v'2-cosEcoct + 2=f d j" m(=)d_]

where coc ==0 for baseband

fd = 10.0 MHz/volt

m(=) = 200 step linear staircase function or 256 step random amplitude

function

The transmitted chirp signal is written as

-- 1

i(t) = "_2 cos[coot 4- _- #t 2]

where

_C = 0 for baseband simulation

/_ == 10.0 MHzJ26 _s

6, Interference Analysis II (Pnlctlcal Modeling and Simulation) 1 7 9
ORIGINAL PAGE iS

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Simulation results in the time and frequency domains are illustrated in the following figures.

Fig. 19 shows the transmitted analog FM-TV signal at baseband. The original message signal for

television is the random function in Fig. 18 with 0.5 volt dc bias. Fig. 20 also illustrates the

corresponding frequency spectrum (power spectral density) of the transmitted signal when the

message signal is random. As another simulation model, the video is modeled by a 200 step linear

staircase function; Fig. 21 shows the same form as Fig. 19 but with this linear step function as its

message signal. Fig. 22 shows thecorresl_onding spectrum. For the transmitted chirp signal, Fig. 23

shows the time waveform and Fig. 24 illustrates _ts spectrum.

The spectral plots, Fig. 20, Fig. 22. and Fig. 24 give the clues for an effective overlay service.

Fig. 22 illustrates the transmitted FM-TV s_gnal spectrum in baseband, with its message signal

modeled as a 200 step linear decreasing stmrcase function, and Fig. 20 shows the same TV signal

spectrum but with a 258 step random amphtude function as its video information. In both cases, the

spectrum is nearly fiat up to approximately 4 2 MHz. However, from 4.2 MHz to 16 MHz, the average

level of the spectrum in the random v,deo message case is much higher than that for the linear

staircase function. Even though neither of these two cases is a real TV spectrum, they can be used

as reference spectra for the design of a ch_rt_ overlay system. The spectrum depicted in Fig. 22 can

be used as a general case of typ0cal "IV _tdeo spectrum, whereas that depicted in Fig. 20 can be

considered a worst case spectrum. The fast decay of the energy in the spectrum in Fig. 22 (linear

staircase message signal) above 42 MHZ sl'lows that this "IV signal can be overlaid by a chirp signal

if the relative power levels of the ct_rl:) and TV s_gnals are set correctly in the transmission stage.

The link analysis in the next section _11 prowde the appropriate value of chirp signal suppression to

prevent serious interference of the _# _qdeo reception, and also gives the specific value of

compression gain required for chirp demodutahon under the power limited situation of typical satellite

links.

As mentioned before, the rne_sa_e _._nal (television video s_gnal) used in the simulation up

to this point has not been processed _ a ;reeml3hasis filter. Preemphasis increases the energy at

higher frequencies in the FM-TV s_ec'-,'_ ,,,n,c_ _ncreases the interference _nto the chirp signal

6. Interference Analysis II (Practical Mo<_e_NI i_l $_mulatlon) 1 8 0
channel.Thesimulationwas repeatedwitha preemphasis filter in the video channel to get results

both in the time and frequency domains and to see the effect of preemphasis filtering.

8. Interferlnce Analysis II (Practical _ =m=lSimulation) 1 S1
I

III

i

E

II

Q

LU

-0_

QDZ

I __

__, _F'-'_U_

Im

II I.

_'I _'O _'O _'O

::L w8

~

'.D

_8

OD

_r8O8

LL/

]E

8,- N8

_D8

O) 8

oo:o°

[A ](1)W

(2 8E

,rod

_R

Z}E

"6

r_@E

F-o

l.l.

Q i

U

Ii

a.

IIM

_>,

gll C

,,( gCC
I I I

I I i

Figure 19. Transmitted FM.TV signal _rne Olot iranclom)

6. Interfertnce Analysis II (Practical Modeling and Simulation)

183
Ol,- 09-

(9PJZ mm I(_-I)SI

0

A

E0

C:

_P

LZ

"6E

f#1

:>- UC@

Ibl_

,{:)

LL

ql'

O0

_o

::I ECC

'10 0A

_u

I:;

Q.vw

"E

l=

,<OUCQ

't:: C
8.00 16.00 24.00

TIME

Figure 21. Transmitted FM-TV signal time plot (linear)

6. Interference Analysis II (Practical Modeling and Simulation) 1 II S
//

sp'z,, l,_sl

N

"r

O

001-

O

ILl

n"h

111 OcF

¢D

L_

"6E2UC@g

e_

0aEcmC0mU

D.Wm

rIIU

I=m
r.=t

k.J

n

Z:

@

O4

¢3

qr

o

O

'It

o

I

@

O4

I

! I

I I I t I

[_=0.3846MHZ/_S, CHOU=O-Z6_S, N=Z048]

1,00 8,00 16,00 24.00 32 .Oil 40.00 48.00 66,00

TIME

_.oo [/,Ls]

Figure 23. Transmitted chirp signal _,_

6. Interference Analysis II (Practical lie.seil_ =_1 $1n_uLIIt|on) I 8 7
CO

N

m

m

h

w

O3

O

O

a

!

T

8

I

mmm

8 /

FREQ, MHz

1 I i

IJ=O.:3846. CHOU=261JS

I

ONLY

18

Figure 24. Frequency spectrum of Figure 23

6. Interference Analysis II (Practical Modalln9 and Simulation) 1 8 8
6.3. Analog FM-television signal modeling II- with preemphasis

In the previous analysis, the models used for the FM-television video signal did not include

preemphasis before modulation of the carrier. However, most practical FM systems employ

preemphasis and deemphasis. Therefore, to get a more practical simulation result, preemphasis

filtering of the analog FM television signal is necessary. Theoretically, a deemphasis filter also must

be included after the television signal demodulation. But in the simulation performed here, especially

the interference analysis at the input of the IF filter, it is not thought to be necessary, because

deemphasis is achieved with a low pass filter which simply eliminates the residual interference spike

after demodulation. The deemphasis filter can be eliminated from the simulation without losing

validity of the simulation result. Fig. 25 illustrates an example of a NTSC FM television modem block

diagram which employs preemphasis and deemphasis, along with typical frequency response curves

of each filter.

6.3.1. Preemphasis Characteristics - NTSC 525 line/simulation

The characteristics of typical preemphasis filters can be found in relevant references.

Janskey and Jeruchim [33] describes the CCIR Pre-Emphasis technique in detail, and Freeman [31]

presents the various preemphasis curves under CCIR Recommendation 405. For more detailed

information see CCIR Report 464 [34}

Fig. 28 illustrates the preemphasls characteristic curves for 405, 525, 625 and 819 line

television systems. Here, in the simulation, we have confined our analysis to the 525 line NTSC

system. Fig. 27 shows the 525 line NTSC preemphasis curve which we used as a model for simulation.

An algorithm was developed by trial and error variation of the parameters of the usual expression for

preemphasis. Fig. 28 shows our s=mulated preemphasis characteristic curve. Although there are

6. Interference Analysis II (Practical Modeling and Simulation)

189
some differences between Rec 405 and the simulated curve, the error is within _ ldB over most of the

frequency range.

6. Intarference Analysis II (Practlc_ _ _ Simulation) 190
; -2  ̧

__ 4

• "6

i. ":O

"+2

fI

I

'_2"J- **he/_

sys,*m / /_n,

._._._ _v,'"_

84._eGan4 tfe4u4ncy (Ml,_l)

_oau_loe Otmo_uk]tOe

F,Jlql,r_ w_'_+lQ_o_141

Sp4clroI OmDIifu_II

al, /

II.|

...... J .41

I

I_.cew_

o_tpuI

I_ II 8,?.'h-**he

* _t 4 S_-kn+\ \

o,-.o'mo_,_zo's; _ _,_o

90'MI4a_4 frequency (l_l)

(t_e-emp_$o$ ct_tocterishc CC,[R I:_:. 4(_)

Figure 25. Typical NTSC modem stT_Cturo ,,,m pree_phesis/ deemphasis

8. Interference Analysis II (Prac_iad _ an4 Simulation) 1 91
10

8

6

O,dlmee,4m_frt_L_'y (MHz)

Figure 2(]. "IV preemphasis character, st_ct,_-25 825. 819 and 405 lines)

6. Interference Analysis II (Practical)4e4me_ll =_1 Simulation) 1 g 2
10

8

6

2

_ =mwQ_mD u

I

i

i

I

10

Figure 27. NTSC 525 line-CCIR Rec.405 _ee_Dr_asJs characteristic

5. Interference Analysis II (Practical Mode.rig -rid Simulation) 1 9 3
ORIGINAL PAGE IS

OF POOR QUALITY

|

8

,i

2,

s

W

'7 t,&a I

i

II/

/

/

/

/

/

i

/

I.I_4

R_I3[I'IMZ_]

/

/ Ill

llll

IIIit

! .Eel I,l_lt

Figure 28. Simulated preemphasis filter for NTSC 525 line "IV system

6. Interference Analysis II (Practical Modeling and Simulation) 1 9 4
6.3.2. Modeling of FM-television video signal - With preemphasis

Basically, there cannot be any differences in the simulation models defined in Section 6.1.A

and 6.1.B, because only a filtering operation has been added before frequency modulation of the

wideband television signal. But for the convenience of simulation, some changes are needed because

of the complex valued form of signal used in the filter process. The active luminance signal duration

was changed from 17 ~ 51 /_s to 18 ~ 50 _.s, which should not cause significant difference in the

spectral domain. The other difference from the model in Section 6.1.B (random amplitude function of

television video) was the number of random amplitude steps. It is necessary to keep the same

number of time or frequency samples throughout the overall simulation procedure, and the filtering

routine was made to accept data in this way. There are 2048 samples over the 84/_s duration of a

single horizontal scanning line; a section lasting 32/_s (18/_s tO 50/_s) which contained 1024 samples,

was selected as having random amplitude. Therefore, instead of selecting and generating 256 random

numbers, 1024 random numbers were generated in this simulation for the preemphasis version.

Fig. 29 shows the video signal with random amplitude before preemphasis, and Fig. 30 shows

its preemphasized version. Because of the gain factor in the preemphasis filter at higher frequencies,

the random portion has enhanced amplitude compared with the non-preemphasized case shown in

Fig. 29. The other signal model was simulated by a staircase function. Fig. 31 illustrates the video

message before preemphasis, and Fig. 32 shows the preemphasis version. In Fig. 32, a transient

spike at 50p.s is caused by the large level shift in the original message signal from -0.SV to 0.5V.

6. Interference Analysis II (Practical Modeling and Simulation) 1 9S
ORIGINAL PAGE IS

OF POOR QUALITY

!

III.llt,[,_lJhl LI.JLll,.JlJt,ll Ill

_=.

..... ill

. _qlqlp,IPl'!'i,"l _ =_

l

E

16 _vm_32 48

M U t_

64

Figure 29. TV message re(t) before preemor_as=s_tJme plot) ;random amplitude

|

6. Interference Analysis II (Practical Modeling and Simulation)

196
ORIGINAL PAGE IS

OF POOR QUALITY

i

Figure 30. TV message re(t) after _.eer,_o_a=,-,ume ;)tot) ;random amplitude

6. Interference Analysis II (Practlaa _ =_1 $1n_ulstlon) 197
ORiG_IAL PAC_ _S

OF POOF_QU,_LJTY

!

!

=!

-.....

'-,,,,

I

Figure 31. "IV message re(t) before _reemohas_s(tlme plot); linear staircase function

6. Interference Analysis II (Practical ModeJin9 and Simulation) 1 9 $
ORIGINAL PAGE B$

OF POOR QUALIYY

Figure 32. "IV message re(t) after preeml_as=s(time plot); linear staircase function

6. Interference Analysis II (Practical Modellng and Simulation) I 91
6.4. Wideband FM television signal simulation I1o-with

preemphasis

The computer simulation was repeated using the same procedures as were described in

Section 6,2, but with the preemphasized v=deo _nformation (message) signals depicted in Fig. 30 and

Fig. 32 of the previous section. Fig. 34 shows the frequency spectrum of the transmitted FM-television

signal for baseband simulation when the video signal is a preemphasized 200 step staircase function.

Fig. 33 illustrates the frequency spectrum of a transmitted FM-television signal for baseband

simulation when the video signal is a preemphasized 1024 step random amplitude function. When

we compare the transmitted wideband FM television spectra which were preemphasized (Fig. 33 and

Fig. 34) with non-preemphasized ones (Fig. 20 and Fig. 22) (i.e., comparing Fig. 20 and Fig. 33 and

comparing Fig. 22 and Fig. 34), we can observe that the average power density level at higher

frequencies in the preemphasized spectra are raised relative to the non-preemphasis cases. The

reason for this can be attributed to the gain factor in the preemphas_s curve at the higher frequencies.

However, the total average power is the same _n all cases because the carrier amplitude of _ is the

same for all signals. In an FM signal, the transmitted power depends solely on the carrier amplitude,

regardless of the modulating signal.

6, Interference Analysis II (Practk:_ Me4e_ _ Slmuimtton) 2 0 0
|!

!

!

!

• I

16 [MHz ]

Figure 33. Transmitted FM-'I'V spectrum , _t)- ramclom, preemphasis version )

6. Interference Analysis II (Practic_ Me4ee_l _ $1mulatlon) 201
I t ' :

i

1 t

i

• I m • • • 4 i •

q IQ

4 | 12

i

g • • I

1 6 f MHz]

Figure 34. Transmitted FM.TV Sl:)_:trur- , ,,_T_- .,_ear. preemphasis version )

6. Interference Analysis II (Practical Nle,,zle4_,g _ Simulation)

202
6.5 Overlay System Link Analysis

In the previous section, 6.2, we have simulated the transmitted television signal and chirp

signal at baseband, and found that a sufficient amount of suppression of the chirp signal permits the

overlay of a chirp signal on top of a television signal, and proper compression gain in the chirp

demodulator can guarantee successful chirp demodulation. The determination of the parameters _n

a given system must be obtained through a link analysis.

The major difference between telev=sion transmission and chirp via a satellite lies in the

power limitation of the earth station in the case of the chirp signal because of the small antenna s_ze

and low transmitter power which are assumed for chirp transmission. The other major difference _s

the receiver bandwidth. Compared to the large IF bandwidth used for FM television, a comparatively

narrow IF bandwidth (about 5 MHz) _s anticipated for chirp. From this point of view, appropriate

selection of the chirp carrier frequency _n the 38 MHZ transponder bandwidth used for television

transmission plays an important role in dectding the carrier to interference ratios (C/I ratio). The final

performance measure of the proposed coded multiple chirp s.s. system is the probability of bit (chip)

error or word (code) miss probability For the FM-TV service, the C/I ratio is a measure of degradation

caused by chirp overlay.

In this analysis, a typical link using the Galaxy II satellite in C band was used as an example

of overtay service. Some key parameter values of "Galaxy It" are listed below.

Galaxy II Parameters [Based on FM.TV Transmission with Chirp Overtay]

Satellite Transponder Parameters

EIRP 34 dBW

G/T .3 to -,-3 dBK-'

6. Interference Analysis II (Practical Modeling and Simulation)

203
Bandwidth 36 MHz

Earth Station Parameters

Antenna Diameter

Aperture Efficiency

System Noise Temperature

FM-TV Receiver IF BW

9.0 m or 2.4 m (for Chirp only)

65

100K ( 120K for chirp )

32 MHz (5.0 MHz for Chirp )

Desired Performance

Analog FM-TV Down Link

Chip Error Probability

C/N > 10 dB

p.< 10-4

The proposed overlay service can be briefly explained here. The high power television signal at

6 GHz is transmitted by a 9 meter antenna and is received by a smaller antenna (2.4 meter in

diameter) via the 4 GHz downlink, At the same time, the chirp signal with relatively low power is

transmitted through a 2.4 meter antenna and received by the 9 meter antenna. The next two tables

shows the FM.television and chirp s.s system link analyses in C-Band.

6.5.1. FM-TV Oveday Link Analysis

Based on the Galaxy II parameters and desired performance crieteria, the link analysis of FM-TV

operating in C-band is listed in following Table 1.

6, Interference Analysis II (Practical Modeling and Simulation) 204
Table 1. FM-TV Overlay Link -- C band

Uplink Parameters

Max Transmitter Power

Antenna Gain ( 9 m )

EIRP

Path Loss ( 6 GHz, Clear Air )

Rain Attenuation

20 dBW (IOOW)

53.2 dB

73.2 dBW

200.2 dB

YdB

Satellite Parameters

Flux Density -85.0 dBW/m z

G/T 3.0 dBK -1

C/N (30.96- Y) dB

Downlink Parameters

EIRP 37.0 dBW

TWTA Output Backoff 3.0 dB

Path Loss ( 4 GHz, Clear Air) 196.3 dB

Rain Attenuation X dB

Rx Antenna Gain ( 2.4 m ) 38.18 dBK-'

G/T 17.28 dBK -1

C/N ( 8.64- X ) dB

6. Interference Analysis II (Practical Modeling and Simulation)

205
6.5.2. Chirp S.S Overlay Link Analysis

As the other phase, chirp spread spectrum overlay link was analyzed as in Table 2. Parameters

in this table were found under the assumption that chirp's transmitted EIRP is lower than FM-TV uplink

EIRP by 38 dB,

6. Interference Analysis II (Practical Modeling and Simulation) 2 0 6
Table 2. Chirp Overlay Link -- C band

Uplink Parameters

Max Transmitter Power

Antenna Gain ( 2.4 m )

EIRP

Path Loss ( 6 GHz, Clear Air )

Rain Attenuation

-4.52 dBW (350mW)

41 7 dB

37 2 dBW

200.2 dB

XX dB

Satellite Parameters

Flux Density

G/T

C/N ( IF BW = 5.0 MHz )

C/N(IF BW = 40KHz)

-121.0 dBW/m z

30 dBK-'

(1 58-XX) dB

(22.8 - XX) dB

Downlink Parameters

EIRP

TWTA Output Backoff

Path Loss ( 4 GHz, Clear Air )

Rain Attenuation

G/T

Chirp Carrier Power

C/N ( IF BW = 5.0 MHz )

C/N (IFBW --- 40KHz)

[1 0 4- (X - XX)] dBW

30d8

19@3 dB

YY dB

29 66 dBK -_

[._48 6 ÷(X-XX)-YY] dBW

-704 dB

'3 94 dB

6. Interference Analysis II (PractlcaJ 04e4e_m_l=rut Simulation) 2 O 7
Probability of Chip Error 1,76 xlO -e

Note.

1. 38 dB attenuation is given under 36 dB compression gain of chirp

2. the probability of Chip error is base on the formula in sec,3.2.4

6. Interference Analysle II (Practical Modelin(j and Simulation)

208
7. Performance Analysis of Overlay Service I-Without

Preemphasis

In the overlay link analysis in previous section, when the chirp signal EIRP is -38 dB below the

FM-televison signal EIRP, the final chip error probability for the chirr) system is around 1.0x10 -4,

assuming 2400 bps( hence corresponding chip rate is around 40K chips/sec using 16 bit codewords

). However, the link analysis clone in previous section does not cons=tier the interference factor when

calculating C/N ratio; it cannot be a sufficient proof that the overlay service is safe from mutual

interference. Calcu!ation of the C/I ratio is needed, so that we can recalculate the final C/N ratio both

in the FM-television demodulator output and in the dechirper output, which will finally give t.he

codeword error(miss) probability.

In the analysis procedure, the signal powers of the television and chirp are calculated first at the

earth station receivers. The FM-telev=s=on s=gnal power and chirp signal power at the receiving earth

station are calculated when each signal works as the interfering signal in the overlay service. Finally

the C/I ratios for both cases are found and plotted for comparison. Because we dealt with two cases,

the preemphasis case and the no-preemphasis case, the analyses will be done separately, then the

final results will be compared.

7. Performance Analysis of Overlay $erv,ct I-.Wlthout Preemph=sls

209
7.1. Signal Power Calculation

A. FM.Television Power

FM-television signal power was set to one watt watt and it can be calculated by

p 1"A .2= _j_')2 1.ow

tv = -E't tv) 2" =

when the transmitted FM-Tv signal is expressed by

t

W(t) = ,J2- cosl'_ct -I- 2_rfd ./" m(_,)d=]

where _= = 0 in the baseband, fd=lO MHz/VoIt, m(¢) is a staircase function or random amplitude

function. In simulation, once the power spectral density of each data point is obtained, the total power

for 2048 samples is the summation of the spectral density over all frequencies, i.e,

n=2048

Ptv= _ Ix(i) 12

L=I

where Ix(i)I = is the power spectral density at the 'rth sample(frequency) point.

The resulting FM-TV signal power calculated by summing spectral density values was:

case1; video signal is 256 step random amplitude function

P_,= 1.00537 W

case2; video signal is 200 step staircase function

P_,=0.97927 W

When we compare the simulated FM-TV power with theoretical power, they are almost same ( within

2.1%).

7. Performance Analysis of Overlay Service I..Without Praemphasis 21 0
B. Chirp Signal Power

Let the transmitted chirp signal be given by

1 2

i(t) = _ cosE_ct + -_-#t ]

then theoretical power is set to one watt:

1 2

Pchirp = "_'(Acl_irp) = 1.0W

By computer simulation, we get the simulated chirp power as

Pcnirp= 0.42213W

The simulated chirp power is 3.7 dB lower than the theoretical value. But, this is expected because

the chirp signal is applied only 26 /_s of the-64_.s scanning period. Thus, 26/64= 0.406 Watt is the

correct value. Because the simulated power is within 44, error, it was assumed that the simulation was

correct.

C _.36 dB Attenuated Chirp Power

In the link analysis, a chirp signal EIRP of -36 dB relative to FM-TV transmitted EIRP was assumed.

Therefore, if we choose a simulated FM-TV power of 1.00537 watt in section A, -36 dB attenuated chirp

power would be 1.00537x10-3s=0.0002528 watt. Figure.35 illustrates this attenuated chirp spectrum; in

this case, relative to the FM-TV signal of which the video signal is a 256 step random amplitude.[ see

Figure.24 for comparison.]

7. PerformanceAnalysisof Overlay Service I.-Without Praemphasis 211
0

O.

N

I

9o

. I

O,J

I

w

U3

8

I0

i

i

r

i

i

.........

A I IZ

FRI_Q , MHz

18

Figure 35. 36 dB attenuated chirp s_gn_ to_:_m

7. Performance Analysis of Overlay 5e_ |-W_:_ I_aemphasis 21 2
7.2. Interference Power Calculation

In section 7.1, FM-TV signal power and chirp power at the earth station receiver were found both in

theory and by computer simulation. As the second step, we calculated the mutual interference power

I, and thus the C/I ratio, when we overlapped the chirp's attenuated spectrum onto the FM-TV signal

transponder bandwidth and vice versa.

7.2.1. Chirp Interference to FM-TV

As mentioned earlier, FM-TV signals use the whole 36 MHz transponder bandwidth, but the overlaying

signal, in this case the chirp signal, uses a relatively narrower bandwidth. We have already fixed the

chirp signal bandwidth at 5.0 MHz in the overlay link analysis. However, because of a restriction of the

FFT routine permitting only 2" number of samples in frequency, a 32 MHz "IV bandwidth was assumed.

Because of this difference in bandwidth, to calculate the chirp interference into the FM-TV channel, it

is necessary to find the power spectral density of chirp signals over a 5 MHz bandwidth. Fortunately

the chirp spectrum has a relatively high level and a fiat portion below 2.5 MHz. Therefore it is not

required to move a 5.0 MHz bandwidth unit to and from the entire 32 MHz (two sided) bandwidth of the

spectral plot. [Note: All the spectral plots are drawn only for positive frequency, and the maximum

frequency in those plots is 16 MHz.]

We already know that the low power chirp signal has a power of 0.00025W relative to the FM-TV

signal of one watt. When we calculated the single sided power of the chirp signal from 0 to 2.5 MHz

bandwidth, we got 0.00012W. therefore the total power in a frequency bandwidth of-2.5 MHz to 2.5

MHz is almost the same as 0.00025W We can conclude that at least 99_ of the power in the chirp signal

is in the -2.5 MHz to 2.5 MHz range

7. Performance Analysis of Overlay Serwce I.-V_thout Preemphasis

213
7.2.1.A FM-Television C/I Ratio

From the result that almost all of chirp power is in the -2.5 MHz to +2.5 MHz band, with a total

power of 0.00025W, there is no difference in C/I ratio if the chirp carrier frequency is located anywhere

in the 32 MHz bandwidth used by the FM television signal. In addition, chirp interference is an all pass

signal in the wideband television IF filter, so interference power from the chirp loses nothing.

Therefore,

C 1.000537[W]

= = 36dB

I 0.00025[W]

7.2.1.B Net ClN of FM-TV under chirp Interference.

By the result of 7.2.1.A, C/I is 36 dB. It is necessary to recalculate the C/N for the TV signal

because the interference adds to thermal noise and affects the C/N. Because the chirp signal power

is 36 dB attenuated relative to the "IV signal, the interference power of the chirp signal is calculated

as -163.12 dBW at the receiver, based on the FM-TV link analysis. If we regard the interference as

another form of noise, the net noise power is -132.76 dBW. Therefore the net C/N under chirp

interference is 8.636 dB. When we compare this C/N with the C/N defined in the link analysis, which

does not consider the interference, there is almost no difference. Therefore we can conclude that ihe

effect of interference on FM-TV is negligible.

7.2.2. FM-Television Interference on Chirp

As the reverse case of section 7.2.1, interference from FM-TV into the chirp channel, the high powered

FM-TV signal is the interference into the chirp channel. Before starting the analysis, the following must

be noted:

7. Performance Analysis of Overlay Service I-.Wlthout Preemphaaia

214
1. Compared to the wide bandwidth of FM-TV (32 MHz assumed), the chirp bandwidth is narrower

at 5 MHz.

.

.

By (1), contrary to the previous case, only a 5.0 MHz bandwidth portion of the FM-TV power

spectrum is interfering with the chirp signal.

By (2), there can be a large difference in interference level depending on the IF center frequency

offset between the chirp and the "IV signal center frequency, within the 32 MHz bandwidth.

7.2.2.A. Chirp CII Rat/o Calculation.

From the result that almost all of chirp power (36 dB attenuated) is located in the -2.5 MHz to

2.5 MHz range, we set the chirp power in a 5.0 MHz bandwidth as C=,,_, = 0.00025W.

The (3/I ratio for the chirp channel depends, in part, on the center frequency used for the chirp

signal relative to the center frequency of the FM-TV signal. Figure 36 showing the spectrum of the

FM-TV signal indicated that the TV signal has a much higher power spectral density close to the

carrier than well away from it. To determine the C/I ratio as a function of frequency offset between the

chirp and TV signal carrier frequencies, we shifted the center frequency for the chirp carrier across

the 32 MHz band of the FM-TV signal in 0.5 MHz steps, and calculated the power from the "IV signal

in a 5 MHz bandwidth centered at the chirp carrier.

Table 3 lists all the necessary information when the FM-TV video signal is modeled by a 256 step

random amplitude function and when the FM-TV video signal is modeled by a 200 step staircase

function. As can be seen from Table 3, when the IF offset frequency approaches the positive maximum

(or negative maximum) (far from the center frequency), the C/I ratio gets better. In addition, when the

FM-TV signal is modeled by a staircase function, the C/I ratio is much better than that of the random

amplitude function. This is expected because the spectrum of the FM-TV signal in this case (staircase

function model) is decaying faster than that of the random amplitude case. Figure 36 and Figure 37

show the overlaid spectral plots of the FM-I'V spectrum modeled by a staircase function (Figure 36)

7. Performance Analysis of Overlay Service I.-W'ithout Preemphasis 21 S
and the FM-TV spectrum modeled by a random amplitude (Figure 37), with a 36 dB attenuated chirp

spectrum. As a graphical representation, the C/I ratio vs IF offset frequency for both cases of FM-TV

interference signal are drawn in Figure 38. Based on Table 3, when the chirp IF center frequency is

located at 12.5 MHz from the center frequency of the FM-TV signal, the corresponding C/I ratio is -5.56

dB. The reason why the best C/I ratio is selected from Table 3, where the message signal (video)

model is a random amplitude function, is that this should be a worst case analysis. Then the net C/N

(EJNo) recalculated for the existence of FM-TV interference, is -9.37 dB. When we compare this value

with the C/N ratio in the earlier link analysis, about 1 dB degradation occurs because of the FM-TV

interference. Finally, if we assume 23 dB compression gain, the C/N ratio after the compression filter

will be + 14dB, the corresponding probability of chip (bit) error will be around 5x10 -6 . In this case, the

data rate is around 25 kbps.

7. Performance Analysis of Overlay Service I-.W1thout Preemphasis 21 6
(D

(D

O.

o° rl,irTI v,

-0 Ir "

i

P_lmmlllili _ ...... ,.................... _.

i

i I

'o 4 l 12 I0

FREO MHz

X FM-TV

CH I RP

Figure 36. Overlapped spectrum of FM.I_ ,rn_t)-I,near stair) and 36 dB attenuated chirp

7. Performance Analysis of Overlay 5e_ I-Wqlt1_out Preemphasis 21 7
x FM-TV

,_, CHIRP

'0 4 8 12 16

FREQ , MHz

Figure 37. Overlapped spectrum of FM.'Pv' ( re(t)- random amplitude ) and 36 dB attenuated chirp

7. Performance Analysis of Overlay Service I..Without Preemphasis 21 8
Table 3. C/I ratio vs IF offset for FM-TV spectrum [No preemphasis] [re(t): 256 step random function

and 200 step linear step function]

256 step random 200 step linear

I" I I I

Leading Trailing Offset 5MHz BW SMHz RW

freq freq freq Tv POWER CH Tv POWER Cil

[,.zI [_.z] [,.,] [wI [d,] [wI [d,]

2.5 -2.5 0.0 0.289 -306 0.2.0 -29.8

3.0 -2.0 0.5 0.287 -30.6 0.239 -29.8

3.5 -1.5 1.0 0.274 -30.3 0.238 -29.8

4.0 -1.0 1.5 0.275 -30.4 0.238 -29.8

4.5 -0.5 2.0 0.253 -30.0 0.242 -29.8

5.0 0.0 2.5 0.414 -32.2 0.387 -31.9

5.5 0.5 3.0 0.459 -32.6 0.471 -32.8

6.0 1.0 3.5 0.426 -32.3 0.449 -32.5

6.5 1.5 4.0 0.407 -32.1 0,422 -32.2

7.0 2.0 4.5 0.375 -.31.8 0.396 -32.0

7.5 2.5 5.0 0.353 -31.5 0.369 -31.7

8.0 3.0 5.5 0.332 -31.2 0.344 -31.4

8,5 3.5 6.0 0.307 -30.9 0.318 -31.0

9.0 4.0 6.5 0.285 -30.5 0.292 -30,6

9.5 4.5 7.0 0,272 -30.4 0.266 -30.3

10.0 5.0 7.5 0.086 -25.4 0.103 -26.1

10.5 5.5 8.0 0.013 -17.4 0.000 0.0

11.0 6.0 8.5 0.010 -16.1 0.0002 4.4

11.5 6.5 9.0 0.007 -14.6 0.0001 6.9

12.0 7.0 9.5 0.005 -13.7 --

12.5 7.5 10.0 0.004 -12.2 m

13.0 8.0 10.5 0.003 -10.8 --

13.5 8.5 11.0 0.002 -9.4 _

7. Performance Analysis of Overlay Service I-.W1thout Preemphasis 219
14.0 9.0 11.5 0.001 -7.4

14.5 9.5 12,0 0.001 -6.6

15.0 10.0 12.5 0.001 -5.6

15.5 10.5 13.0 0.001 -5.7

16.0 11.0 13.5 0.001 -5.9

m m

u

7. Performance Analysis of Overlay Service I--Without Preemphesis 2 2 0
8

8

6

tM

U

F

8

0

_r

I

8

IO .00 3.00 6.00 9.00 12.00 IS .EEl

FREQ [ MHZ ]

X RANDOM

• LINEAR

Figure 38. C/I ratio vs IF offset frequency (message are linear stair and random amplitude )

7. Performance Analysis of Overlay Service I.-VVlthout Preemphasis 2 21
8. Performance Analysis of Overlay Service II- With

Preemphasis

In chapter 7, we have analyzed the interference effects in an overlay service, but without preemphasis

filtering at the input video signal to the FM-TV modulator. In this section, the same procedures as in

section 4 are repeated for the case when preemphasis is considered. Based on the spectrum and time

plots of the transmitted wideband signal _n section 64, the chirp signal is also attenuated by 36 dB

relative to the FM-TV signal spectrum Ftgure 39 and Figure 40 show the overlapped spectra of chirp

and FM-TV signals. For simplicity and to avoid repetitive explanation, only the result of the

calculations are described here.

8.1. Signal Power

1. FM-TV signal power (by simulatsonl

• <Case 1 > when video stgnal ,s mo(leled by random amplitude, the TV power is 1.00507 W

8. Performance Analysis of Overlay Ser,n¢_ II. With I_'Qemphasis 2 2 2
| W II I • • • It I IN !1 I_ • • •

!0

12

4 8

lie

16 [ MHz]

Figure 39. Overlapped spectrum of FM.T'V rpreemphasis) and 36 dB attenuated chirp(random)

8. Performance Analysis of Overlay Serv_(:e II. Preemphasia Case. 223
!

!

W

i,

!

.,,i

,.i

=,

illi

- :_=_-

I

i

i !

m=

NM

WWl

m

16 [MHz]

Figure 40. Overlapped spectrum of FM-'I'V (preempehsis) and 36 dB attenuated chirp(linear stair)

8. Performance Analysis of Overlay Service II- Preemphasis Case. 2 2 4
• < Case 2 > when video signal is modeled by staircase function, the TV power is 1.02763 W

. Chirp signal power is same as before because preemphasis is not applied to the chirp signal.

However after 36 dB attenuation, it turn out to be 0.00025 W. Because the simulated TV power is

within 1% error from the theoretical value of 1.0W, we need no normalization of the FM-TV

spectrum. In addition, the 36 dB attenuated chirp power can be used as before.

8.2 Interference of Chirp to FM-TV

By using the same procedures in section 6.2.2.A, almost all the chirp power is in the -2.5 MHz to

2.5 MHz bandwidth. Therefore, 1=0.00025 W and C=1.00537 W,and the C/I ratio will be 36 dB. In

addition, the net C/N ratio of FM-TV under chirp interference is the same as before because the C/I

ratio is the same as before.

8.3 Interference of FM-TV into chirp channel.

By following the same procedure as in section 6-2-3, Table 4 can be obtained for the case when

the FM-TV video signal is modeled by random amplitude function and the FM-TV video signal is

modeled by the staircase function, respectively. Figure 41 illustrates the C/I ratios versus IF offset

frequencies for the different video signal models. Figure 42 depicts the four cases of C/I ratio vs IF

offset frequencies. One pair is for the cases when preemphasis is used and the other pair is for the

cases when no preemphasis is used. In Figure 42, the C/I ratio still maintains the trend that C/I ratio

gets better as the TV carrier frequency goes away from the chirp IF center frequency. In Figure 42,

we can observe the effect of preemphasis. The use of preemphasis in the FM-TV video signal,

8. Performance Analysis of Overlay Service II- With Preemphasis 225
regardlessofthemodulationapplied,degradestheC/Iratioapproximately15-20dBingeneral.Figure

43 shows the calculated chip error rate when the C/N of the chirp signal is found for the case of FM-TV

signal interference. From the result for FM-TV interference, the final C/N ratio of the chirp signal is

found to be -20 dB before compression. If 23 dB compression gain is provided, the probability of chip

error is around 10-1 . When this is compared to the case for no preemphasis in chapter 6, about 105

times more error can occur in the preemphasis case. However, the compression gain can be

arbitrarily increased up to 10000 using recent advanced technology, although the cost may be a

critical factor. Around 5-10 dB increase in processing gain can be used to compensate for this

degradation caused by preemphasis filtering.

8. Performance Analysis of Overlay Service II- Wtth Preemphasis 2 2 6
Table4.Cll ratiovs IFoffsetfor FM.TVspectrum[WithPreemphasis][m(t):256steprandomfunction

and200steplinearstepfunction]

256 step random 200 step linear

I I I I

Leading Trailing Offset 5MHz BW 5MHz BW

freq freq freq Tv POWER CII TV POWER CII

[.,.] [,,.] [,..] ] [do] ] [,=,o3

2.5 -2.5 0.0 0.788 -34.5 1.001 -36.1

3.0 -2.0 0.5 0.788 -34.5 1,001 -36,1

3.5 -1.5 1.0 0.551 -33.4 0.752 -34.8

4.0 -1.0 1.5 0.523 -332 0.682 -34.4

4.5 -0.5 2.0 0.497 -33.0 0.605 -33.8

5.0 0.0 2.5 0.442 -32.5 0.511 -33.1

5.5 0,5 3.0 0.393 -32.0 0.417 -32.2

6.0 1.0 3.5 0.365 -31.8 0.339 -31.1

6.5 1.5 4.0 0.332 -31.2 0.266 -30.3

7.0 2.0 4.5 0.087 -25.4 0.011 -16.1

7.5 2.5 5.0 0.069 -245 0.007 -14.8

8.0 3.0 5.5 0.061 -23.9 0.005 -13.4

8.5 3.5 6.0 0.052 -23.1 0.005 -13.0

9.0 4.0 6.5 0.043 -22.3 0.003 -11 0

9.5 4.5 7.0 0.041 -22.1 0,001 -7.4

10.0 5.0 7.5 0.035 -21 5 0.0005 -3.5

10.5 5.5 8.0 0.033 -21 1 0,0003 -1.5

11.0 6.0 8.5 0 031 -20 9 0.0003 -1.5

11.5 6.5 9.0 0 028 -20 5 0.0004 -1.7

12.0 7.0 9.5 0025 -20 1 0.0004 -1.7

12.5 7.5 10.0 0 024 -19 8 0.0003 -1.2

13.0 8.0 10.5 0 022 -19 4 0.0003 -1 2

13.5 8.5 11.0 0 021 .19 2 00002 -0.1

8. Performance Analysis of Overlay Sarv_ It. lihllempNiIde Case. 2 2 7
I

!II

00

IIII

iiiI

c_ c_ _- _-

@WuMW

i-nEo

a.BmQMwc@u

t:E0@

0.
o

U

8

8

,¢

I

O

O

!

0.00 3.00 6.00 9.00

FREQ[MHZ]

12.00 1S.00

x RFCqOOM

LINERR

Figure 41. C/I ratio vs IF offset frequencylpreemphasis case)

8. Performance Analysis of Overlay Service II. Preemphasis Case. 2 2 9
8

o

i

8

!

LJ

8

IO .00

FREQ [MHZ ]_

X

RA_PR

LI NEARPR

RANDOM

LINEAR

Figure 42. C/I ratio vs IF offset frequency(preemphasis case and no preemphasis case)

8. Performance Analysis of Overlay Service II. Preemphasis Case. 230
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9. Conclusions

Multiple-coded chirp has been proposed as a valuable spread spectrum communication technique for

low data rate services. It avoids the d_fficult and lengthy synchronization process of all direct

sequence spread spectrum systems, making _t especially valuable for low duty-cycle applications.

Chirp can be used to overlay an FM-TV channel provided the chirp center frequency is offset from

the TV carrier. Study of a satellite channel carrying a "rV signals shows that two chirp signals with data

rate up to 25 kbps could be overlaid in a 38 MHz transponder without significant mutual interference.

Performance estimates for a VSAT earth station operating at C-band show that a 2.4 meter antenna

and 300 mW transmitter could send a 24 kbps sfgnal to a large central earth station over an occupied

channel.

9. Conclusions 23 2
10. References

1. J.R. Klauder, et al. "The Theory and Design of Chirp Radars," Bell System Technical Journal, vol

29, No. 4, July 1960.

.

V. Ramanan, "An Asynchronous Multiple Access Scheme for Satellite Communications," Ph.D.

Dissertation, Department of Electronic and Electrical Engineering, University of Birmingham,

England, June 1983.

3. R.S. Berkowitz, "Modern Radar," John Wiley, New York, NY, 1965.

4. C.E. Cook, M. Bernfetd, "Radar Signals", Academic Press, New York, NY, 1967.

5. H.O. Ramp, E. R. Wingrove, "Principles of Pulse Compression," IRE Trans. Military Electronics,

vol. MIL-5, April 1961, pp 109-116

6. R.C. Dixon, "Spread Spectrum Systems," John Wiley, New York, NY, 1976.

7. M.R. Winkler, "Chirp Signals for Communications," IEEE WESCON Convention Record, 1962.

10. References 233
8. E.K. Holland-Moritx, J. C. Dute, and D. R. Brundage, "Swept Frequency Modulation, Proc. Nation.

Electron. Conf. vol. 22, 1966, pp. 469-474.

9. A.J. Berni, W. D. Gregg, "On the Utility of Chirp Modulation for Digital Signaling," IEEE Trans.

Comm., vol. COM-21, June 1973, pp. 748-751.

10. W. Hirt, S. Pasupathy, "Continuous Phase Chirp (CPC) Signals for Binary Data Communications."

IEEE Trans. Comm. vol. COM-29, June 1981, pp. 836-858.

11. D. S. Dayton, "Coming to Grips with Multipath Ghosts," Electronics, vol. 40, Nov. 1967, pp 104-108.

12. G.F. Gott, J. P. Newsome, "HF Data Transmission Using Chirp Signals," Proc. Inst. Elec. Eng., vol

118, pp. 1162-1166, Spet 1971.

13. G.W. Barnes, D. Hirst, D. J. James, "Chirp Modulation System in Aeronautical Satellite," AGARD

Conf. Proc., No. 87, Avionics in Spacecraft, Royal Aircraft Establishment, Rome, Italy, 1971.

14. Max Kowatsch, J. T. Lafferl, "A Spread-Spectrum Concept Combining Chirp Modulation and

Pseudonoise Coding," IEEE Trans., vol. COM-30, No. 10, Oct. 1983, pp. 1122-1142.

15. P. W. Baier, R. Simons, H. Waibel, "Chirp-PN-PSK Signale als Spread - Spectrum - Signalformen

geringer Dopplerempfindlichkeit and grosset Signalformvielhalt," NTZ Archiv. vol. 3, pp. 29-33.

Feb. 1981.

18. M. Kowatsch, F. J. Siefert. J. Lafferl, "Comments on Transmission System Using Pseudonoise

Modulation of Linear Chirps." IEEE Trans. Aerosp. Electron. System, vol. AES-17, pp. 300-303, Mar

1981.

17. R. Gold, "Characteristic L_near Sequences and Their Coset Functions," J. Soc. Ind. Appl. Math,

May 1965.

10. References

234
18. T.Kasami, "Weight Distribution Formula for Some Class of Cyclic Codes," University of Illinois,

Urbana, Rept. R-265, Apr. 1966.

19. C. E. Cook, "Pulse Compression - Key to More Efficient Radar Transmission," Proc. of IRE, vol

48, pp. 310-316, Mar. 1960.

20. B.A. Carlson, "Communications Systems, McGraw Hill," 1975

21. M. Cohn, W. L. Eastman, "A Class of Balanced Binary Sequence with Optimal Autocorrelation

Properties" IEEE Trans. Inform. Theory, vol. IT-23, pp.38-42, 1977.

22. D. V. Sarwate, M. B. Pursley, "Crosscorrelation Properties of Pseudorandom and Related

Sequences," Proc. of IEEE, vol. 68m No. 5, pp. 593-619, May 1980.

23. D.V. Sarwate, "Bound on cross-correlation and autocorrelation of sequences," IEEE Trans. Inform

The__ vol. IT-25, pp 720-724, 1979.

24. S. Stein, "Unified Analysis of Certain Coherent and Non-Coherent Binary Communications

Systems," IEEE Trans. Inform. Theor?, vol. IT-IO, pp. 43-51, Jan. 1964

25. C.W. Helstrom, "Resolution of Signals in White Gaussian Noise," Proc. IRE Tvol. 43, pp. 1111-1118.

Sept. 1955.

26. L.W. Couchll, "Digital and Analog Communication Systems," pp. 362-364, MacMillan, New York.

NY 1983.

27. T. Pratt, C. W. Bostian, "Satellite Communications," John Wiley, Inc., New York, NY 1986. p!0

241-242.

28. R. M. Gagliardi, "Satellite Communications," Lifetime Learning Publications, Belmont, CA. 1983.

pp. 248.

10. References

235
29. K. Feher, "Digital Communications," Prentice-Hall, NJ, 1983, pp. 375-379.

30. F. G. Stremler, "Introduction to Communication Systems," Addison- Wesley, 1982, pp.338-341.

31. R. L. Freeman, "Telecommunication Transmission Handbook," John Wiley, Inc., Boston NY 1981,

pp.194.

32. M. A. Maggenti, "Spread Spectrum Multiple Access and Overlay Service," M.S. Thesis, VPI&SU,

Mar 1987.

33. D. M. dansky, M. C. Jeruchim, "Communication Satellite in Geostationary Orbit," Artech House,

1983, pp.482-485.

34. CCIR: XlV'th Plenary Assembly, Kyoto, Japan, vol.4, Report 464 on "Pre-emphasis Characteristics

for Frequency Modulation Systems for FDM Telephony in the Fixed Satellite Service," Genova,

1978.

10. References

236