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Advanced Space Communications

Architecture Study

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Volume 2 - Technical Report

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NASA CR 179592

IJI _. --L 4 N N'/

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Prepared for

National Aeronautics and Space Administration

Lewis Research Center

Cleveland, OH 44135

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Prepared by

TRW Eiectromc Systems Group

One Space Park

Redondo Beach, CA Y0278

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Advanced Space Communications

Architecture Study

Volume 2 - Technical Report

NASA CR 179592

March 1987

Preparedfor

National Aeronautics and SpaceAdministration

Lewis ResearchCenter

Cleveland,OH44135

Prepared by

TRW Electronic Systems Group

One Space Park

Redondo Beach, CA 90278
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Advanced Space Communications

Architecture Study

Volume 2- Technical Report

NASA CR 179592

March 1987

Prepared by Prepared for

| TRW Electronic Systems Group NASA Lewis Research Center

One Space Park 21000 BrookparkRoad

| RedondoBeach, CA 90278 Cleveland,OH 44135

!

Approved: !

' _Y-_ c-__,_'__,__- -

M.G. Davis, Jr.

I Principal Investigator Manager, Electronic Development Operation

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TABLE OF CONTENTS

Introduction

System Description and Analysis

2.1 System Concept

2.2 Link Design

2.3 SatelliteReceive Antenna

2.4 Bulk Demodulators

2.5 SatelliteTransmit Antenna

2.6 Baseband Multiplexer

Satellite Receive Antenna Design

3. I MBA with Multiple Reflectors

3.2 MBA with MultipleLenses

Satellite Transmit Antenna Design

4.1 Phased-Array-Fed Reflector Antenna

4.2 Active-Aperture Phased Array

4.3 MBA with Beam Switching Matrix

4.4 Comparison of Antenna Configurations

Bulk Demodulator Technology

5.1 SAW Demodulators

5.2 FFT Demodulators

5.3 Conclusions

Baseband Multiplexer Technology

6.I High-Speed SerialTime-Domain Multiplexers

6.2 Frequency-Domain Multiplexers

6.3 Memory-Based Multiplexers

Transmitter and Receiver Technology

Satellite Weight and Cost

8.1 Payload Weight

8.2 SatelliteWeight

8.3 SatelliteCost

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pa_a_

1-1

2-1

2-1

2-4

2-9

2-12

2-17

2-26

3-I

3-3

3-8

4-I

4-I

4-14

4-25

4-32

5-I

5-2

5-12

5-22

6-I

6-3

6-6

6-11

7-I

8-I

8-I

8-14

8-19
,

10.

11.

TABLE OF CONTENT5 - continued

User Terminal Design and Cost

9.I Terminal Description

9.2 Key Design Issues

9.3 Terminal Cost

5ubscriber Charge

Conclusions

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9-I

9-2

9-8

9-II

10-I

11-I

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6-I

8-I

8-2

8-3

8-4

8-5

8-6

8-7-

8-8

8-9

8-10

8-11

9-I

9-2

9-3

10-I

10-2

10-3

10-4

10-5

LIST OF TABLES

Uplink Power Budget (29.5GHz)

Downlink Power Budget (19.7GHz)

EIRP for Array-Fed Dual Reflector Antenna

EIRP for Active-Aperture Phased Array

Typical MultiplexerRequirements

Payload Weight and Power Summary

ReceiverlDemultiplexer Weight and Power Estimate

Modulator Weight and Power Estimate

Array-Fed Dual Reflector Antenna Weight and Power

Estimate

Satellite Power Requirements

Satellite Weight vs. Eclipse Capability

Satellite Weight Budget

Thermal Control Subsystem Weight

Satellite Cost Summary

Payload Cost Summary

Spacecraft Bus Cost 5ummary

LNA Implementation Tradeoff

Unit Cost Estimates

User Terminal Cost Estimate

Traffic and User-Terminal Profiles

Assumptions for Service Charge Calculation

Program Expenditure Profilefor Minimum Number of

Terminals

Required Revenue ($000 per Channel-Year)

Subscriber Service Charge (¢ per Channel-Minute)

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4-11

4-18

6-2

8-3

8-6

8-12

8-13

8-15

8-16

8-17

8-18

8-20

8-22

8-23

9-9

9-12

9-13

10-2

I0-6

10-9

10-14

10-15
LIST OFFIGURES

I-I Satellite Links

2-I

2-2

2-3

2-4

2-5

2-6

2-7

2-8

2-9

2-10

3-I

3-2

3-3

4-I

4-2

4-3

4-4

4-5

4-6

4-7

4-8

5-I

5-2

5-3

5-4

5-5

5-6

5-7

5-8

5-9

i-3

SatellitePayload FunctionalBlock Diagram 2-3

UplinkBeam Pattern 2- I I

SingleFFT-Bulk-Demodulator Power Requirements 2-14

Bulk Demodulator Power Requirements for Composite 2-16

IO00-MHz Bandwidth

Phased-Array-Fed Gregorian Reflector Antenna 2-19

RF Power Requirements forMagnification Factor of 2 2-2i

RF Power Requirements for Magnification Factor of 3 2-22

Main Reflector Diameter for Array-Fed Dual Reflector 2-23

Antenna

Two-Antenna Transmit Configuration 2-24

Onboard Data Storage Versus Downlink-Beam Switch Time 2-28

Uplink Beam Pattern

Offset-Fed Reflector Antenna Geometry

Multiple-Lens Antenna Geometry

Array-Fed Dual Reflector Antenna System

Phased Array Transmit Modules

Radiating Horn Edge-of-Coverage Gain

Hexagonal Array EIRP

Linear Array Equivalents to Hexagonal Array

MBA with Beam Switching Matrix

Triangular Beam Pattern

Beam Interlacing with Four-Aperture Antenna System

SAW Demodulator

Compressive Receiver Block Diagram

Reflexive Array Compressor (RAC) Operation

High Degree of Correlation Between Model and Hardware

Recursive FFT Using VHSIC I Chips

Pipelined FFT Block Diagram

Image Signals Due to Phase and Amplitude Mismatch

Effect of Amplitude and Phase Mismatch on Signal-toImage Ratio (SIR)

Presumming Calculations

Page iv

3-2

3-4

3-9

4-2

4-3

4-16

4-21

4-22

4-26

4-27

4-29

5-3

5-5

5-6

5-8

5-14

5-16

5-17

5-19

5-21

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6-I

6-2

6-3

6-4

6-5

6-6

7-I

7-2

7-3

7-4

7-5

7-6

7-7

8-I

8-2

8-3

9-I

9-2

9-3

10-I

10-2

10-3

10-4

10-5

10-6

LIST OF FIGURES- continued

Bulk Demodulator and Buffer for Time-Domain Multiplexer 6-5

Time-Domain Multiplexer SerialInterconnection 6-7

Time-Domain Multiplexer ParallelInterconnection 6-8

Frequency-Domain Multiplexer 6-9

Address-Pointer Memory-Based Multiplexer 6-13

(Wire-OR Version)

Raster Output Memory-Based Multiplexer 6-15

Device Power Level Projections

Device Efficiency Projections

Solid-State Amplifier Design Example

Device Noise Figure Projections

Device Gain Projections

TRW HEMT Performance Projections

Low-Noise Amplifier Design Example

Satellite Payload Functional Block Diagram

Receiver/Demultiplexer Block Diagram

Modulator Block Diagram

User Terminal Block Diagram

User Terminal Configuration

IMPATT Amplifier Design

7-2

7-3

7-5

7-7

7-8

7-9

7-10

8-2

8-4

8-II

9-3

9-4

9-10

Page v

Satellite Development and Manufacturing Schedule 10-8

Program Expenditures for Minimum Number of Terminals 10- I 1

Program Expenditures for Twice Minimum Number of 1O- 12

Terminals

Program Expenditures for Three Times Minimum Number ! 0-13

of Terminals

Nondiscounted Cash-Flow Profile for Minimum Number of 10-18

Terminals

Discounted Cash-Flow Profile for Minimum Number of 10-19

Terminals
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1. INTRODUCTION

Until recently, it was generally believed that trunking traffic would

domlnate polnt-to-polnt satelllte communlcatlons. Because of the rapid

progress In fiber optlcs communlcatlons, however, It Is now clear that

trunklng traffic wlll, for the most part, be carrled terrestrlally rather

than by satelllte. In the future, therefore, polnt-to-polnt satelllte

communlcatlons wlll be domlnated by thln-route appllcatlons Involving

flxed or moblle termlnals.

NASA's Advanced Communlcatlons Technology Satelllte (ACTS)

program was concelved durlng the perlod when trunklng trafflc was the

domlnant point-to-point satellite mode. Accordingly, ACTS is intended to

advance technology development In both trunklng systems and customer

premlse service (CPS). The latter servlce Is Intended to provide satellite

volce and/or data llnks dlrectly between end users, thereby obviating

constructlon of terrestrlal talls between earth station and end user. Wlth

the anticipated demlse of satelllte trunklng, the CPS aspect of ACTS is the

more relevant to future satelllte communlcatlons.

Thls reportpresents the resultsof a 12-month study to determine

satellite system architectures that are most suitable for CPS

communications. As with ACTS, the focus Is on 30120-GHz (Ka-band)

technology. However, the system architecturesexamined are equally

applicableto operationat 14/II GHz (Ku-band).Because largenumbers of

Page I-I
users (i.e., 10,000 or more) are intended to be served, emphasis is placed

on system designs that permlt low user terminal costs.

The satellite links In the type of system under consideration are

shown In Figure 1-1. Complete "single-hop" lnterconnectivity, whlch ls

needed for voice communications, ls provided between any pair of system

users. To establish such a connection, a user requests elther a one-way

channel or a two-way circuit, as needed, from the master control station.

These requests are transmitted over an orderwire channel, typically

making use of a random access protocol. When the circuit is no longer

needed, It ls restored to the pool of clrcults available for reassignment.

Minimization of terminal costs generally Implies low values of peak

EIRP. It ls also Important to llmlt the terminal size, since real estate and

Installation costs might otherwise outweigh termlnai equipment costs.

The terminal antenna diameter ls llmlted to 1.2m in most of continental

United States (CONUS), although l.Sm antennas are permitted In

heavier-rainfall areas to malntain system availability. A

frequency-division multiple access (FDMA) upllnk signal format ls

selected because It permits narrow-bandwidth transmission, which tends

to mlnlmlze EIRPrequirements.

Single-hopinterconnectlvltyrequiresthat the satelliteact llkea

switchboard in the sky,wlth accessabllityto each individualterminal

transmission,The acceptedmeans of providingthlsaccessablllty(andthe

^n,, ,,,,_ArnnIA,,,-,,l ,.,,,_,,,,y_ I_,, ,,yACT.,, C_I ie,.,use of a tirn_,-rliui_inn ....................... multipleaccess (TFIHAI

transmissionformat.The use of FDMA, on the otherhand,heretofore

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II! / / IIIIII \ \ \ \

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would have required a separate satellitedemodulator for each terminal

transmission. However, the prospect now exists for the development of

"bulk" demodulators of sufficiently low power consumption to be

attractive for satellite use. A bulk demodulator a_cepts as input a

composite FDMA signal and provides a tlme-dlvision multiplexed (TDM)

pulse stream in which the demodulated bits of the various input carriers

are Interleaved.Thus, the demodulator output has a format similar to that

of the uplInktransmissions In a TDMA system. A single bulk demodulator

might simultaneously process 100 FDMA carriers, thereby providing a

100:1 reduction In the number of demodulators that would otherwise be

required.

In contrast to the upllnk,downllnk transmission Is accomplished by

means of a small number of wideband, TDM carriers. A baseband processor

buffers the relatlvely large number of demodulator outputs and

multiplexes them onto the downlIn.kcarriers.

The satellite transmit antenna is designed to have a relatively high

galn (l.e., small beamwldth), to mlnlmlze the transmlt power

requlrements. In addltlon, the small spot slze produced by the antenna

permits the Introductlon of frequency reuse, whlch expands the effectlve

system bandwidth. A separate antenna beam is generated for each

downlink carrier. Each beam is "hopped" among the downlink spots,

dwelling on each spot long enough to transmit a predetermined number of

bits from each 64-kbps upllnk carrier Intended for one of the termlnals

_A_,II.I_t_ t-k,_ ,,',._,._t- k,-,.._,.,l-..,,,,_,

VV II.,11111 _,11_ _l, JIJ_, I,Jt,JUl II, J_l )/*,

Page I- 4

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A complete system descriptionand a number of key system tradeoffs

are presentedIn Section2. Detailsof thesatellitereceiveand transmit

antennadesignsare presentedir_Sections3 and 4, respectively.The bulk

demodulatortechnologyIsexamined In Section5, the baseband processor

technologyIn Section6,and transmitter/recelvertechnologyat 30120 GHz

InSection7.

Sections 8 to II are concerned wlth the financial aspects of system

implementation and operation. The weight of each payload element,

together wlth a satelllte subsystem welght budget, Is presented In Section

8. Nonrecurrlng and recurring satelllte cost estlmates, derived from the

welght estlmates, are also provlded In Section 8. A user terminal design

and cost estlmate are glven In Section 9. Sectlon I0 derives the user

service charge needed to provlde a speclfled rate of return on Invested

capital, based on an assumed traffic profile. The study conclusions are

stated In Section I I.

Page I- 5
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2. SYSTEH DESCRIPTION AND ANALYSIS

Because the ground segment is a major system cost driver, the first

step in defining a system concept is to select an appropriate set of user

terminal parameters. By means of the link equations, corresponding

satellite receive antenna gain and EIRP requirements are then obtained to

satisfy the system performance objectives. Subject to the derived

satellite requirements, a number of satellite subsystem trades are

performed to arrive at a complete system description.

2.1. System Concept

Each terminal is provided the capability to transmit a single

narrowband digital carrier. The minimum or standard data rate is taken as

64 kbps. The carriers transmitted by the various terminals are

distinguished by frequency and/or polarization. Thus, the uplink signal set

comprises a large number of FDMA carriers spread over the 500-MHz

Ka-band fixed-satellite allocation not shared with the fixed (terrestrial)

service. Since dual polarization is used, the equivalent composite

bandwidth is 1000 MHz.

With present-day digital voice representations, the 84-kbps terminal

transmission can support two voice channels. With the higher-speed

digital circuitry expected by the end of the decade, 18-kbps digital voice

should replace the current 32-kbps voice standard. Each terminal will

then be capable of transmitting four voice channels. Hore generally, a

terminal could support various combinations of voice and data totaling. 64

Page 2- I
kbps. The constituents of the 64-kbps transmission need not be directed

at the same terminal. With QPSK transmission employed on the uplink (see

Section 2.2), the in-phase and quadrature components are readily

separated in the satellite and retransmitted to different terminals.

A functional block diagram of the satellite payload is shown in Figure

2-I. The satellite receive antenna produces a number of fixed area beams

which provide complete CONUS coverage. The carriers received on a given

beam are divided into groups of standard size, each of which is input to a

bulk demodulator. The latter converts a set of FDMA signals, contiguous in

frequency, to a single TDM pulse stream in which pulses corresponding to

an individual input signal are interleaved with those of all other inputs.

The demodulator bandwidth is chosen narrow enough so that the traffic

handling capability of each beam can be matched to the traffic offered

through assignment of an appropriate number of demodulators. Allowing a

certain number of demodulators to be switchable between beams leads to

accommodation of an epr/or/unknown or time-varying traffic pattern.

The bulk demodulator outputs are fed to a baseband multiplexer which

buffers and sorts the data bits according to the downlink beam for which

they are intended. The number of downlink beams is considerably smaller

than the number of spots (as measured by the antenna beamwidth) needed

to cover CONUS. Each downlink beam is therefore programmed to scan over

a portion or all of CONUS in a repetitive pattern. The data bits assigned to

a beam for transmission during one complete scan are organized according

to the downlink spot for which they are intended. In this way, no spot is

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Page 2-3
visited more than once per scan by any of the beams.

As depicted in Figure 2-I, each downlink beam is formed by

power-dividing the corresponding carrier and feeding the resulting signals

to modules associated with the radiating elements. Each module consists

of an amplifier and a variable phase shifter. Each radiating element is

excited by the combined output of several modules, one for each downlink

beam. The set of radiating elements can either form a direct-radiating

array or be used as an array feed for a reflector type of antenna.

2.2. Link Design

The user terminals are equipped with 1.2m antennas. This size is

typical of very-small-aperture-terminal (VSAT) design. Each terminal

that transmits the standard 64-kbps carrier has a 2W transmitter. This

power levelisabout the maximum achievable at reasonable cost. Both the

terminal and the satellite are equipped with HEMT LNAs. The

corresponding receiver noise figures are projected to be 2.0 dB and 3.0 dB,

respectively.

Considerable uncertainty exists concerning the rain margin required

to achieve a specified link availability. The margins selected are intended

to correspond to an uplink or downlink availability of 0.995 in all but the

southeast portion of CONUS (Rain Zone E), as reported in Reference 2-I.

Accordingly, no end-to-end transmission path has an availability of less

than o.gg. In fact, most composite paths have considerably higher

availability. An antenna diameter of 1.8m is required to extend 0.995 link

availability to Rain Zone E.

Page 2-4

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A higher link availability than 0.995 was not selected because of the

impact on the satellite design. Improving the link availability to 0.9975

(i.e., halving the link outage) requires uplink and downlink margins of about

15 dB and 6 dB, respectively, compared with 8 dB and 4 dB for a link

availability of 0.995.

To maximize system capacity, QPSK transmission is used on the

uplink. The data is differentially encoded to simplify the demodulation

process on the satellite. On the downlink, the choice of modulation is

dictated by the need to minimize the cost, and hence the complexity, of the

receive portion of the user terminals. To this end, differential data

encoding is essential. BPSK is selected in place of QPSK because of the

larger Eb/N o requirement associated with differential QPSK. The larger

carrier bandwidth associated with BPSK transmission necessitates a

degree of spatial frequency reuse, in addition to polarization diversity.

This requirement imposes an operational constraint on the downlink-beam

scanning patterns, if excessive interbeam interference is to be avoided.

To provide the requisite link quality, assumed to correspond to a BER

of 10-7, end-to-end error correction coding is employed. Use of

soft-decision decoding on the ground would be largely negated by the

demodulation process on the satellite. Therefore, hard-decision decoding

is used instead.

Because of the relatively small EIRP and G/T of a 1.2m terminal, it is

desirable to obtain as much coding gain as possible. On the other hand, a

lower code rate leads to a larger bandwidth per carrier and therefore

Page 2-5
smaller system capacity.

rate-3/4 coding.

These opposing factors lead to a choice of

Uplink and downlink power budgets are given in Tables 2-1 and 2-2.

Each link is nominally designed for a decoded BER of 10-7 (a fictitious

quantity in the case of the uplink). The end-to-end error rate of the

undecoded transmissions is twice that of either the uplink or the downlink

considered separately. Therefore, the BER of the decoded transmissions

will in fact be somewhat greater than 10-7 .

The required values of Eb/N o in Tables 2-1 and 2-2 include an

implementation margin (i.e., departure from theoretical) of 3 dB. In

addition, the Eb/N o penalty due to differential data encoding is 2.3 dB on

the uplink (for QPSK) and 0.6 dB on the downlink (for BPSK).

The required uplink signal quality is achieved by proper design of the

satellite receive antenna, while the downlink signal quality depends on

proper selection of the satellite EIRP. Accordingly, the receive antenna

gain must be at least 33.3 dB throughout CONUS. For the selected downlink

data rate of 180 Mbps per beam, the satellite EIRP must be at least 63.1

dBW.

The choice of 180 Mbps for the downlink data rate is governed by the

desire to accommodate the downlink data on four carriers. As will be seen

in Section 4, this downlink data partitioning leads to use of two identical

Page 2-6

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TABLE 2-I. UPLINK POWER BUDGET (29.5 GHz)

Transmitter power (2W), dBW

Line loss,dB

Antenna gain (1.2m),dB

Pointing loss,dB

EIRP, dBW

Path loss,dB

Atmospheric loss,dB

Rain loss (.995 except Zone E),dB

SIC antenna gain (edge of beam) plus linelosses,dB

Received carrierpower, dBW

System noise temperature, dB-K

Boltzmann's constant, dB (WIK-Hz)

No,dB (WIHz)

Bit rate (64 kbps),dB (sec-1)

EblN o,dB

Page 2-7

3.0

-1.0

49.2

-1.0

50.2

-213.7

-0.6

-8.0

33.3

- 138.8

27.6

-228.6

-201.0

48.I

14.1
TABLE 2-2. DOWNLINK POWER BUDGET (19.7 GHz)

EIRP, dBW

Path loss, dB

Atmospheric loss, dB

Rain loss, dB

E/S antenna gain, dB

Pointing loss, dB

Line loss, dB

Received carrier power, dBW

System noise temperature, dB-K

Boltzmann's constant, dB (W/K-Hz)

No, dB (W/Hz)

Bit rate ( 180 Mbps), dB (sec- 1)

Eb/N o, dB

Page 2-8

63.1

-210.2

-0.8

-4.0

45.8

-0.8

-10

- 107.9

-202.9

82.6

12.4

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transmit antennas, each radiating a pair of carriers on opposite

polarization. The downlink data rate is therefore one-fourth of the total

satellite throughput.

The satellite throughput is determined by uplink considerations. Each

uplink carrier supports a data rate of 64 kbps. With QPSK transmission

and rate-3/4 coding, the transmitted symbol rate is 42.67 ksps. For

purposes of bulk demodulation, the carrier separation must be twice the

transmitted symbol rate, or 85.33 kHz. The effective system bandwidth of

l O00 MHz accommodates 11,718 such carriers or, equivalently, a

composite data rate of 750 Mbps. Allowing 30 Mbps for system overhead

leaves 720 Mbps as the system throughput rate. The downlink data rate is

therefore !80 Mbps per beam.

Two 180-Mbps carriers are transmitted on each polarization. With

rate-3/4 coding and BPSK modulation, each carrier supports a

transmission rate of 240 Msps. With a bandwidth-to-symbol rate ratio of

(say) 1.5, each downlink carrier occupies 360 MHz. With a 500-MHz

allocation on each polarization, the two carriers clearly have overlapping

spectra. As observed earlier, this overlap places constraints on the

downlink-beam scanning patterns.

2.3. Satellite Receive Antenna

From Table 2-1, the satellite receive antenna must provide a

minimum gain over CONUS of 33.3 riB. This minimum gain determines the

required number of uplink beams. For a given angular coverage, the gain at

beam edge is maximized if it is chosen 4.3 dB less than the peak gain.

Page 2-9
Therefore, the peak gain must be 37.6 dB. This gain provides beam

coverage (i.e., 4.3-riB beamwidth) of 2.56 degrees. Approximately eight

such beams, forming the pattern shown in Figure 2-2, are needed for

CONUScoverage.

The portion of the frequency .allocation, and correspondingly the

number of bulk demodulators, assigned to each uplink beam is proportional

to the traffic offered by user terminals located within the beam coverage

area. The frequency band assigned to a given beam may be confined to a

single polarization or divided between the two polarizations. The first

alternative requires only a single port per feed horn (rather than a

separate port for each polarization) and leads to a simpler repeater

configuration. For this reason, it has been chosen for the baseline

satellite configuration.

It should be pointed out, however, that confining each beam to a

single polarization has the effect of dividing the total available bandwidth

of 1000 MHz into two distinct 500-MHz allocations. Consequently, the

system saturates once the combined traffic in the beams assigned to

either polarization requires a bandwidth of 500 MHz. The satellite

throughput at which saturation occurs can be increased, however, by

taking advantage of the beam overlap in Figure 2-2. Users located in

overlap areas can be assigned to either of two satellite beams. If the two

beams in a particular case have opposite polarization, proper user/beam

assignments can help balance the traffic division between the two

polarizations.

Page 2- i 0

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Two possible multiple-beam antennas (MBA) are discussed in Section

3, one based on offset-fed reflectors and the other on multiple lenses. The

former has been chosen for the baseline design. The need for multiple

reflectors arises fromthe relationship, in the case of a single reflector,

between the feed horn aperture and the separation between feed horn

centers needed to generate the beam pattern in Figure 2-2. Proper

reflector illumination requires a feed horn aperture which is too large to

allow the feed horns to be suitably positioned for the desired beam

pattern. Consequently, multiple reflectors are employed. Three reflectors

suffice. Two of the reflectors support three beams, while the third

reflector supports the remaining two beams. Beams generated by the same

reflector are given a common numbering in Figure 2-2. The diameter of"

each reflector is 14 in.

To achieve an uplink availability of (say) 0.9975, rather than 0.995,

between 30 and 40 satellite beams would be required. This would greatly

increase the complexity of the satellite payload and therefore was not

considered.

2.4. Bulk Demodulators

The power requirements of a set of bulk demodulators spanning a

fixed composite bandwidth (1000 MHz in this case) depend on the

single-demodulator bandwidth. Based on this relationship, which is

developed below, a baseline value of demodulator bandwidth is chosen.

Fixing the demodulator bandwidth (and hence the number of demodulators)

establishes the connectivity requirements for the baseband processor in

Figure 2-1.

Page 2- 12

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The power required by a digital bulk demodulator is shown in Figure

2-3 as a function of the number of 64-kbps channels per demodulator, for

different technologies and modes of system operation. A nominal channel

spacing of 100 kHz is assumed. Each demodulator consists of an A/D

converter followed by an FFT unit. The FFT channel capacity is typically

available in blocks which differ by powers of 2. These capacity

increments result in FFT power-requirement discontinuities at certain

demodulator bandwidths. The preferred operating points, in terms of

power consumption, correspond to demodulator bandwidths at which the

FFT is fully occupied. A maximum of about 100 channels can be passed

through a 128-channel FFT when allowance is made for anti-aliasing filter

roll-off. There is no resulting loss of bandwidth utilization, however, as

two such FFTs can process a pair of adjacent 100-channel frequency bands.

Two FFT power consumption curves are shown in Figure 2-3, one for

synchronous terminal operation and the other for asynchronous terminals.

The latter alternative refers to a system in which the terminals are

allowed to be time-offset (but frequency synchronous) with respect to

each other. This approach would simplify terminal design by eliminating

some of the acquisition timing requirements. The impact on FFT power,

however, is quite dramatic for large FFTs because of the need for higher

sampling rates (2 to 3 times higher) and FFT overlapping and combining

operations. In addition, any windowing (for sidelobe reduction) must be

performed as a convolution after the transform, increasing total power.

Page 2- 13
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Synchronous time operation, on the other hand, eliminates the need

for such high-rate sampling and extra FFTs. Time-synchronous operation

also considerably simplifies the multiplexing operation prior to downlink

transmission, since data bit-time relations are fixed rather than variable.

For these reasons, only synchronous operation of bulk demodulators is

considered.

The A/D converter consumes considerably less power than the FFT,

the amount depending on the choice between CHOS and bipolar technology.

Like the FFT, bipolar A/D devices exhibit discontinuous power consumption

as a function of demodulator bandwidth. By contrast, CHOS A/D power

consumption increases smoothly with demodulator bandwith.

The total power requirement for a demodulator complement sufficient

to span 1000 MHz of bandwidth is shown in Figure 2-4. Curves are drawn

for FFT demodulators with both synchronous and asynchronous terminal

operation, as well as for SAW demodulators. All three curves incorporate

the projected CHOS A/D power requirements from Figure 2-3. Power

requirements are based on VLSI and wafer-scale signal processor

projections for the next five years. The SAW curve is based on power

consumption of 25 watts per I00 channels. This figure is based on

experience with the MILSTAR SAW demodulators and represents some

power improvement in the A/D area. SAW power requirements tend to be

independent of the number of channels, whereas A/D and detector power

consumption tend to follow curves similar to the FFT A/D. No significant

power reductions due to improvements in SAW technology are expected in

the near future.

Page 2- 15
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the curves in Figure 2-4 indicatea region, in the neighborhood of

50-200 channels per FFT demodulator for synchronous operation, over

which the required power tends to be minimized. The corresponding

demodulator bandwidth is5-20 MHz. A totalof 50-200 such demodulators

are required for a composite signalbandwidth of I000 MHz.

A baseline value of l O0 bulk demodulators, each of l O-MHz bandwidth

and supporting !12 channels, will be adopted for the baseline system.

This choice results in a high degree of parallelism (i.e., !12 channels

demodulated at a time), while permitting the demodulator capability to be

matched to the traffic in each uplink beam to an accuracy of better than 1

percent of satellite capacity.

2.5. Satellite Transmit Antenna

The satellite transmit antenna is required to generate four scanning

spot beams, each supporting a 180-Mbps carrier and producing an EIRP of

63.1 dB (at an antenna gain 4.3 dB below the peak value). Three different

antenna configurations are considered in Section 4: a paraboloid reflector

illuminated by a phased array using a Gregorian subreflector, an

active-aperture phased array, and a switched-beam MBA. The latter

configuration requires the use of multiple high-power (i.e., l OOW) TWTAs.

A significant DC power saving results from the high DC/RF conversion

efficiency of TWTAs relative to that of solid-state devices. However, the

higher reliability of solid-state amplifiers, together with the graceful

degradation resulting from amplifier failure in a phased array, makes the

array-based configurations more attractive than the MBA for this

application. The array-fed reflector antenna, which is depicted in Figure

Page 2- 17
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2-5, is preferred to the direct radiating array because of the reduced array

dimension.

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The beamwidth of the array-fed reflector antenna is determined by

the main reflector dimension, D !. The array dimension, D2, is less than D 1

by the magnification factor, M. A larger value of M results in greater

maximum scan loss, for a given coverage requirement. A value of I"1equal

to 3 provides a good compromise between the dual objectives of

minimizing both array size and scan loss.

An array of given dimension is simplified by a reduction in the number

of elements. The maximum permissible spacing between elements is

dictated by grating lobe considerations. Generation of grating lobes can

produce interbeam interference, as well as a reduction in main lobe gain.

An element spacing that avoids generation of grating lobes within the

coverage area, without degrading the main beam gain significantly, is

identified in Section 4.

For any fixed element spacing, the key tradeoff in the array design is

expressed by:

EIRP ~ PTOT NE = PE NE2 I

where NE is the number of elements, PE is the RF power per element, and

PTOT is the total RF power. The validity of the initial proportionality can

be verified by noting that, for a fixed element spacing, NE is proportional

to the array area, and therefore, for fixed I"1,to the main reflector area and

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gain.

Page 2- 18
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PE and PTOT are shown as a function of NE in Figures 2-6 and 2-7, for

M equal to 2 and 3, respectively. The curves correspond to transmission of

a 180-Mbps carrier with EIRP of 63 dB, and are drawn for an element

spacing, d, approximately equal to the wavelength, _.. At this element

spacing, there is little loss of power to grating lobes. Complete

elimination of grating lobes would require an element spacing of 0.48 ;L For

M=2, and 0.32 ;k for M=3. The main reflector size corresponding to the

curves in Figures 2-6 and 2-7 is shown in Figure 2-8.

For M=3, the baseline value, Figure 2-7 shows the RF power per

180-Mbps carrier for a l O00-element antenna to be 105W. The total RF

power for four such carriers is 420 W. The latter figure depends only on

the 720-Mbps satellite throughput and not on the number of downIink

carriers. The corresponding RF power per element for a single carrier is

105 mW. The main reflector diameter is 64 in., according to Figure 2-8.

The selected method of implementing four simultaneously scanning

beams is shown in Figure 2-g. Following power division, each carrier is

passed through a phase-shifter/amplifier combination for each radiating

element. With N elements and four beams, a total of 4N phase shifters and

4N amplifiers are required. Following amplification, the carriers are

combined in pairs. The carriers in each pair are given orthogonal circular

polarization by passage through an orthomode transducer (OMT) prior to

excitation of the radiating element. Two separate antennas are employed,

one for each pair of carriers.

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Use of a separate amplifier for each carrier element avoids

generation of intermodulation products, which reduce useful output power

and generally add to the overall background noise level. In addition, the

OMT results in essentially lossless carrier combining following

amplification. Therefore, this configuration minimizes the DC power

requirement for a given configuration of radiating elements. The obvious

disadvantage is the need for two distinct antennas.

The data in Section 4 can be used to derive an appropriate antenna

configuration, under the condition that the absence of grating lobes is

required only within the coverage area (i.e., over CONUS). For an array of

hexagonal shape, a minimum element spacing of 1.84 _. suffices. Each

array has 331 radiating elements, and the main reflector diameter is 69

in. The half-power beamwidth is 0.75 degree. Beam coverage, however, is

defined by the 4.3-dB beamwidth. About 60 spots are required for

complete CONUScoverage.

The RF power per carrier with this configuration is 97W. Of equal

importance, the RF power per carrier element (i.e., per amplifier) is 300

roW. This is a modest power level, even for current 20-GHz technology.

The DCIRF efficiency of a single-stage, 20-GHz GaAs amplifier

operated at saturation isprojected to be in the neighborhood of 35 percent

in 1990. The achievable efficiencyof a multistage amplifier, including

the effect of power conversion, will probably be no greater than 25

percent. The latterfigure implies a DC power requirement on the order of

1600W for the amplifiers associated with the four downIink carriers.

Page 2-25
A reduction in DC power can be achieved through an increase in

antenna gain (i.e., larger main reflector), which, for fixed values of M and

minimum element spacing, implies a larger number of radiating elements.

Doubling the number of elements, for example, would halve the DC power

requirement. An assessment of this tradeoff depends on the resulting

satellite weight and cost, and is therefore deferred until Section 8, where

these topics are discussed.

The above results are valid for a downlink availability of 0.995. For a

fixed number of elements, a downlink availability of 0.9975 would

increase the DC power requirement by 2.3 dB. Alternatively, if the DC

power is held fixed, the number of elements must be increased by the same

factor. If either of these alternatives were chosen, the minimum

end-to-end link availability would only be increased from o.9g to 0.9925,

because the uplink availability is only guaranteed to be 0.995.

2.6. Baseband Multiplexer

The baseband multiplexer combines the I00 bulk-demodulator outputs

into four downlink data-streams. Three different multiplexer

configurations are discussed in Section 6. The preferred candidate is

termed a memory-based multiplexer; associated with each demodulator is

a memory that can hold up to 2xlO 4 data bits. The total memory

requirement is therefore 2xlO 6 bits for the l O0-demodulator

complement.

From a system standpoint,a key parameter is the number of bits from

Page2-26

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each 64-kbps uplink- channel that are stored by the multiplexer and

subsequently transmitted as a unit on the downlink. At one extreme, each

uplink quadriphase symbol could be treated as an independent entity.

However, this would require unreasonably rapid hopping of the downlink

beams. The transmission time for each uplink symbol is 23.44 i_s. Suppose

that each of the four downlink beams is required to transmit to 30 of the

60 spots covering CONUS during this interval. In addition, assume that the

time allowed to switch between beam positions is 2 percent of the

average dwell time. The required beam switching time for separate

transmission of each downlink symbol would then be only 16 ns.

The time allowed for beam switching increases in proportion to the

number of uplink symbols per channel stored prior to downlink

transmission. This relationship is displayed in Figure 2-10 for different

numbers of spots/beam covered during a single scan. Beam switching

times of 100 I_S or greater are quite feasible. Therefore, at least five

bits/channel should be stored and transmitted as a unit on the downlink,

for the 30 spots/beam case. This minimum number implies that 112 kbits

of data are buffered by the multiplexer at any time. The permissible beam

switching time can be increased to 400 ns by increasing the onboard data

storage to 560 kbits.

The minimum number of spots/beam is influenced by the uplink and

downlink signal characteristics. Suppose that the user terminals are

required to transmit and receive on opposite polarization. If only a single

polarization is used in each uplink beam, roughly half the downlink spots

will contain terminals that transmit on one polarization, and the other

Page 2-27
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half will contain terminals that transmit on the opposite polarization In

this case, the two downlink beams with the same polarization need only

cover half the 60 spots between them; a single beam must therefore cover

15 spots. On the other hand, if both polarizations are permitted in each

uplink beam, each downlink spot must be visited by at least one of the

beams on each polarization. In this situation, each downlink beam must

cover 30 spots.

Reference:

2-1. "30/20 GHz Demonstration System", TRW Final Report - Volume II,

Documentation No. 36565-018, 29 May 1981

Page 2-2g
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3. SATELLITE RECEIVE ANTENNA DESIGN

As derived in the uplink power budget of Table 2-1, the satellite

receive antenna must provide a gain of 33.3 dB as measured at the receiver

input. For a beam of given angular coverage, the gain at the beam edge is

maximized if it is chosen to be 4.3 dB less than the peak gain. Therefore,

the required peak gain is 37.6 + Ls dB, where Ls is the loss between the

feed horn output port and the LNA input port. A value of Ls = 0.2 dB will be

assumed; therefore, the peak gain must be 37.8 dB.

The corresponding antenna diameter is given by

O=O.l_)4-GT-fi (3-I)

where ;L is the wavelength, (3 is the peak gain, and tI is the antenna

efficiency. For 11= 0.5, D = 34.9 ;L = 13.9 in. The half-power beamwidth,

e3, for a horn-fed reflector can be approximated by

03 = 75 _/D (3-2)

The 4.3-dB beamwidth, e4.3, is given by

e4. 3 = 1.2 83 (3-3)

and, finally, the angle between neighboring beam centers (Figure 3-1 ), eb,

Page 3-1
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is given by

eb = 0.86 04. 3 (3-4)

it follows that 83 = 2.15", e4. 3 = 2.56", and eb = 2.22".

The eight-beam configuration in Figure 3-I includes a rectangular

area measuring 3.5 eb x 1.44 8b. With eb = 2.22",this rectangle is 7.8" x

3.2" and therefore includes the boundaries of CONUS, which measures

approximately 7" x 3". This eight-beam configurationwill be taken as the

baseline uplink antenna pattern.

Two candidate multiple-beam antennas (MBA) are considered, one

based on offset-fed reflectors and the other on multipl.e lenses. From a

system viewpoint, the frequency band allocated to each beam may be

confined to a single polarization or it may be divided between the two

polarizations. In the first case, the feed horns (for either antenna

configuration) have only a single port; in the latter case, a separate port is

required for each polarization. If both polarizations are used in a single

beam, the lens antenna may be preferred because of its superior

polarization properties.

3.I. MBA with MultipleReflectors

This antenna consists of three offset-fed paraboloid reflectors.

geometry for each reflectoris shown inFigure 3-2. The angle,Of,

The

Page 3-3
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subtended by the reflector is given by

ef = tan- l[4tr/(4-E;2)] (3-5)

_vhere _ is the ratio of the reflector diameter, D, to the focal length, f. A

conventional design will be assumed, in the sense that f = D. This leads to

a value of Of equal to 53".

Two of the reflectors are fed by three horns, and the third by a pair of

horns. Beam assignments for the three reflectors are shown in Figure

2-2. Beams generated by the same reflector have an angular separation of

1.5 eb.

The need for three reflectors arises from the relationship between

the feed horn aperture (d I), the spacing between feed horns (Sf), and the

angular separation between beams generated by a common aperture. The

parameter d I is chosen equal to the smaller of Sf or the aperture size that

provides the desired reflector illumination. It can be shown that a

radiation pattern with a -I0 dB intensity (I e) at the edge of the reflector

(referred to the illumination intensity at the center of the reflector)

maximizes the aperture efficiency.

produces the best compromise

underillumination of the reflector.

The reason is that a -10 dB taper

between spillover loss and

(This optimization is a "broad"

function of Ie because, for Ie = -6 dB or -20 dB, the aperture efficiency is

only reduced by 0.5 dB.) The l O-dB beamwidth of a circular-aperture horn

Page 3-5
antenna is given approximately by

O 10 _ 120 _.IdI (3-6)

Substituting elo = ef = 53" in (3-6) and solving for dll_.results in

d I/_" _ 2.26 (3-7)

The value of Sf required to produce a pair of beams with angular

separation e is given by

Sf = f[tan (Fbde)] (3-8)

where Fbd is the beam deviation factor, which typically has a value of I.I.

If the eight-beam antenna pattern were produced by a single reflector, 8

would be equal to eb = 2.22" and Sf would be equal to 0.59 in. or 1.48 _.

The feed horn aperture would be limited to the latter value, rather than

2.26 _ as expressed by (3-7). The resulting spillover loss, with D = 34.9 _,

= 13.9 in., would reduce the antenna gain by 1.7 dB relative to the assumed

50-percent efficiency.

With the three-reflector configuration, on the other hand, the beam

separation for each reflector considered separately is 1.5 eb = 3.33 °. From

(3-8), the corresponding feed horn separation is 2.22 _ = 0.89 in. Thus, d 1,

as weii as Sf, can be chosen equal to _'_' 2_. _'- ,_, u_, z._,z ,_,_ results in a :_-"_

Page 3-6

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illumination taper and the previously assumed 50-percent antenna

efficiency.

The polarization isolation that must be maintained when both

polarizations are used in a single beam depends on the disparity in

received signal strength from different terminals. For signals of equal

magnitude, polarization isolation of 20 dB would result in a l-dB

reduction in overall carrier-to-noise ratio. (A 14-dB carrier

power-to-thermal noise ratio is assumed.) However, the combination of

EIRP variations and rain attenuation can lead to a lO-dB disparity in

received signal strength. Therefore, 30-dB polarization isolation is

required.

An antenna such as the multiple-reflector configuration, or the

multiple-lens configuration described in the next section, has

crosspolarized sidelobes that attain their maximum value at an angle 0c

from the copolarized beam axis. Usually, 8c = 83/2. For linearly polarized

transmission, the crosspolarized sidelobes comprise four "beams" equally

spaced on a circle of radius ec and displaced from the direction of

polarization by 45". With a center-fed antenna, the crosspolarized lobes

are approximately equal in magnitude. With an offset-fed reflector, two

lobes are stronger than the other two and are usually stronger than those

produced by a center-fed antenna. With f/D = 1, the crosspolarized lobes

are about 25 dB below the copolarized beam peak. With careful design and

fabrication, it may be possible to reduce the crosspolarized lobes to -30

dB with respect to the peak of the copolarized beam. With circular

Page 3-7
polarization, the four distinct lobes become a continuous circular lobe at

an angle 83/2 from the copolarized beam axis.

Note that, even if each beam is restricted to a single polarization, the

crosspolarized sidelobes, which are at an angle 03/2 from the copolarized

beam axis, may interfere with transmissions in an adjacent beam. To

avoid this possibility, it is necessary to impose an operational constraint

that precludes frequency reuse in adjacent beams.

3.2. MBA with Multiple Lenses

The rationale that led to the need for multiple reflectors applies to

the lens configuration as well. Consequently, an antenna efficiency of 50

percent may be assumed with a three-lens system. Such an antenna is

shown schematically in Figure 3.3. The lens diameter, D, and the focal

length, f, are again both equal to 13.9 in. The feed-horn spacing and the

diameter of the horn aperture are both 0.89 in.

The lens is designed to convert spherical waves radiated by the feed

horn into plane waves propagating away from the lens aperture. A

first-order design is obtained by choosing the lens thickness so that all

rays, emanating from the focal point and terminating in the aperture plane,

have the same electrical length; the lens thickness, t, is given by

t = [1 -(cos Q/cos Qrn)]/( 1-Jccos $) (3-9)

where $ is the angle indicated in Figure 3-3, 20rn is the angle subtended by

Page 3-8

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Page 3-9
the lens, and cis the dielectric constant of the lens. Using D = f = 13.9 in.,

the thickness of a rexolite lens (c = 2.4) varies from 3.0 in. at the center to

zero at its outer edge.

Lens antennas must use a slow-wave medium to convert the incident

spherical wave to a plane wave. The slow-wave medium has a

characteristic impedance different from that of air or free space. This

impedance mismatch introduces reflected waves and reduces the antenna

gain. If a quarter-wave matching layer is placed over the entrance and

exit surfaces of the lens, this mismatch loss can be reduced from about

1.0 dB to less than O.! dB. Assuming that a matching surface is used, the

lens antenna gain, half-power beamwidth, and scan properties will be

essentially the same as those of the reflector system described

previously.

Crosspolarized radiation of a lens antenna, as well as for a paraboloid

reflector, depends strongly on the f/D ratio. Larger f/D and center-fed

antenna systems have lower response to crosspolarized incident fields.

With the lens being center-fed and the reflector offset-fed, the lens

antenna will have less coupling between orthogonally polarized ports.

The weight of the lens antenna combination is about four times that

of the reflector configuration. In absolute terms, however, the lens

antennas weigh a total of only 25 lb.

Page 3- 10

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4. SATELLITE TRANSMIT ANTENNA DESIGN

The sateilite transmit antenna is required to generate four scanning

spot beams, each supporting a data rate of 180 Mbps and providing an EIRP

of 63.1 dBW. Three different antenna configurations are considered for

this purpose: a paraboloid reflector illuminated by a phased array using a

Gregorian subreflector, an active-aperture phased array, and a

switched-beam MBA. Performance characteristics and estimated size of

these candidate systems are presented below.

4.1. Phased-Array-Fed Reflector Antenna

A single scanning beam can be produced by a phased array illuminating

a Gregorian configuration of dual confocal paraboloid reflectors as shown

in Figure'4-1. Conceptually, the phased array launches a plane wave

toward the smaller paraboloid subreflector. The subreflector intercepts

this energy and converges it to a "point" near, or on, its focal point. This

energy, in turn, is intercepted by the larger (main) paraboloid, which

transforms it into a narrow beam. If the phased array is excited with a

uniform phase distribution, the subreflector converges the intercepted

energy at its focus. Since both reflectors have the same focal point, the

axis of the secondary beam produced by the main paraboloid is coincident

with the common focal axis of the paraboloids. If, instead, the phase

distribution varies linearly with distance from a point on the edge of the

array, the beam axis is scanned off the focal axis.

The array is shown schematically in Figure 4-2a. An input carrier

Page 4- !
Figure 4=1. Array-Fed Dual Reflector Antenna System

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signal at frequency fi is modulated and divided equally among N phase

shifters which, in turn, excite N modules. Each module consists of an

SSPA, a horn radiator, and associated circuitry. The N modules are

assembled to form an N-element array antenna.

The circuit shown in Figure 4-2b has two input carrier signals, f l and

f2- Each carrier signal is divided equally among its own set of phase

shifters. The output of each set of phase shifters drives an associated set

of SSPAs to produce two independent excitation phase distributions, _1

and 0"2, across the array aperture. This, in turn, produces two separate

beams, one for f I and a second for f2-

Antenna Dimensions. The antenna gain is determined by the dimension

of the main reflector. For a fixed EIRP, therefore, the antenna size (and,

as will be seen, the size of the array) can be traded off against RF power.

For purposes of illustration, the RF power per carrier is assumed to be

I OOW. It can be shown that the antenna gain at the edge of a beam of

specified angular coverage is maximized when the peak gain is 4.3 dB

greater than the edge gain. Allowing 0.5 dB for scan loss, the required

peak gain, G, is equal to 63.1 - 20 + 4.3 + 0.5 = 48 dBi.

The required half-power beamwidth, 83, can be estimated using the

approximate relationship,

e3 = _r3-_OO/G (4-1)

Page 4-4

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Substituting G = 63096 (i.e., 48 dBi) leads to the result that e 3 must be

0.75". The antenna gain at an angle e from boresight is given by

G(e) = - 12(8183)2 dB (4-2)

Setting G(e) = -4.3 dB and solving (4-2) for e indicates that the 4.3-dB

beamwidth is 1.283 = 0.9". Consequently, the beam coverage area is 0.9" in

diameter. The O.9-deg beams are overlapping, such that all points of

interest lie within either one or two beams.

The half-power beamwidth of a circular aperture can be estimated

from

e3 = kZ/D (4-3)

where D is the aperture diameter, _. is the wavelength, and k is

approximately 70 for the nearly uniform array illumination with sharp

edge taper. Solving for D/;I gives

D/;L = 70/83 (4-4)

At f = 19.7 GHz, _ = 0.598 in. and, from (4-4), D = 56 in. when 83 = 0.75 °

Hence, the area of the reflector projected parallel to the focal axis will be

56 in. in diameter. The focal length of the paraboloid can vary over a wide

range; however, choosing f = D = 56 in. represents a reasonable

compromise between physical size and scan loss•

Page 4-5
The subreflector is also an offset section of a paraboloid. Its focal

axis and focal point must coincide with the focal axis and focal point of

the main reflector. The projection of the subreflector parallel to the focal

axis is a circle with diameter d. The focal length f' of the subreflector

will be assumed equal to d. The focal length f' can be less than d; however,

it cannot be larger than d. Referring to Figure 4-1, the surface of the

reflectors is given by

p = 2f/(l + cos e) (4-5)

for the main reflector and

p' = 2f'(i + cos e') (4-6)

for the subreflector.

Magnification of the feed array by the Gregorian optics is the primary

advantage of using the array-fed, dual-reflector system instead of a

planar phased array. Assume that the feed array diameter equals d,

although it can be smaller. The dual-reflector system produces an image

of the feed array as indicated. The image has a diameter equal to D, as

long as the feed array diameter is 2d. (If the array diameter is less than d,

the image diameter will, for main beam analysis, be less than D.) The

array diameter is decreased by the magnification factor, M, where

M = Did (4-7)

Page 4-6

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The array diameter is also a function of the number of elements, N, in the

array and the element aperture diameter, do. It is important to note that

radiation patterns and gain can be analytically determined using the image

of the feed array in_ead of integrating currents on the reflector surface.

Number of Elements. The feed array should be filled, rather not

thinned, to cause the angle eg between a grating lobe and the beam axis to

be as large as possible. Choosing a smaller value of do will also cause eg

to be larger. Distributing the elements on a hexagonal grid minimizes the

maximum spacing between adjacent elements and also maximizes eg. It is

important to maximize eg because this reduces the gain of the grating

lobe, thereby increasing the main lobe gain. (This aspect of array design,

which is the same for an active-aperture phased array, is discussed in

Section 4-2.) Consequently, the feed array is assumed to consist of N

elements on a hexagonal grid. Each element has a circular aperture with

diameter do. Adjacent element spacing is uniformly equal to do.

The radiation patterns and gain of the array-fed reflector system will

be established by reference to the corresponding properties of the image

array shown in Figure 4-1 The gain of an array antenna, Ga, is

approximated by

Ga = NGel (4-8)

where Gel is the gain of an array element. Assuming the element has a

uniformly illuminated circular aperture (a good approximation for a horn

Page 4-7
antenna), Gel is given by

Gel(O)= 4(_doM/_,)2[J 1(u)/u]2 (4-9)

where

u = 11Mdosin_/_. (4-10)

and_ is the angle between a ray to the far-zone observation point and the

focal axis. Jl(U) is the Bessel function of order 1 with argument u. The

magnification factor, M, converts the diameter of the array element to the

diameter of its image in the image array.

Ga,as given by (4-8) to (4-10), is the gain at the peak of a beam; the

gain Ge at the edgeof beam coverage is 4.3 dB less. Hence,

Ge(_)= 1.49N(llMdo/_.)2[Jl(U)/U]2 (4-I 1)

The factor Jl(U)/U reduces Ge(_) as _) is increased. The maximum

reduction, Z_Ge, occurs when the beam is scanned to the edge of CONUS.

From geosynchronous orbit, CONUS subtends an approximately rectangular

area measuring 7" x 3 °. Increasing N reduces z_Ge and, as stated before,

increases eg.

Page 4-8

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The satellite EIRP in each of the four scanning beams is given by

EIRP = 1.49N2Pe(_Mdo/;L)2[J l(U)/U]2 (4-12)

where Pe is the RF power per element. To use (4-12) to relate EIIqP, N, and

Pe, it is first necessary to express do in terms of N. The number of

elernents in a hexagonal array varies in accordance with

J

N = 1 + _6i = 1 + 3J(J+l)

i=l

(4-13)

where NO = 2J + 1 is the number of elements along a major diagonal of the

array. However, No is also equal to D/Md o. Equating the two expressions

for No, solving for J, and substituting the result into (4-13) yields

N = 1 + 3 Md° - 1

(4-14)

Practical values of M are limited by the associated scan loss. It can

be shown that the scan loss, Ls, is given approximately by

Ls = 10 log [1 - 4(M+ 1)(f'/d)(_/_)] dB (4-15)

where _ is in radians With f'/d

Ls = 10 log [1 - 078(M + 1)]dB.

= 1 and 0 = 006 radian (ie, _ =35°),

A value of M equal to 3, for example,

Page 4-9
produces a scan loss of i.6 dB. For a selected value of M, the associated

scan loss must be subtracted from the EIRP computed from (4-12) to

obtain the EIRP actually realized.

The EIRP resulting from various combinations of M, Pe, and N is shown

in Table 4-1 for a main reflector diameter of 56 in. For a hexagonal array,

N varies in discrete steps according to the selected value of J in (4-13).

It is seen, for example, that for M = 3 and Pe -- 0.SW, a 331-element array

is needed to produce the required EIRP of 63.1 dBW.

The required RF power per beam ( 165W in this case) can be reduced by

choosing a larger main reflector, with a corresponding increase in the size

of the array. If the element size is held fixed (thereby maintaining the

grating lobe properties), the number of elements, N, increases as the

square of the reflector diameter. The RF power per beam varies inversely

with N, while Pe varies inversely with N2. The larger reflector implies a

larger number of downlink beam positions (i.e., spots) for CONUS coverage.

Increasing M decreases the dimension of the subreflector

proportionately. It also increases the scan loss. For M = 5, D = 56 in., and

N = 331, the element size and the interelement spacing approach the width

of a standard section of rectangular waveguide. Smaller interelement

spacing could lead to difficulty in packaging the amplifiers, phase

shifters, connecting waveguide, etc. Review of the data in Table 4-1

indicates that M = 3 is probably the best choice. Similarity between the

array-fed reflector system and the active-aperture array permits other

Page 4- 10

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Table 4-1. EIRP for Array-Fed Dual Reflector Antenna

Frequency = 19.7 GHz Wavelength= .598

Diameter of Main Reflector= 56 Inches

Maximum Scan Angle= 3.5 Degrees

Inches

Focal Length = 56 Inches

Magnification Factor= 3 Scan Loss=-l.619 dE Power Per Element=

Number of Diameter Loss Due EIRP

Elements of Element To Element (Including all)

(Inches) Factor(dB) Losses} (dBW}

91 1.697 -3.065 55.711

127 1.436 -2.155 58.066

169 1.244 -1.601: 59.859

217 1.098 -1.237 61.306

271 0.982 -0.986 62.522

331 0.889 -0.804 63.572

397 0.812 -0.668 64.496

.5 Watt

Magnification Factor= 3

Number of

Elements

91

127

169

217

271

331

397

Scan Loss=-1.619 dB Power Per Element=

Diameter Loss Due EIRP

of Element To Element (Including a11)

(Inches) Factor(dB) Losses} (dBW)

1.697 -3.065 58.722

1.436 -2.155 61.076

1.244 -1.601 62.869

1.098 -1.237 64.317

0.982 -0.986 65.532

0.889 -0.804 66.582

0.812 -0.668 67.506

1 Watt

Magnification Factor= 4 Scan Loss=-2.139 dB Power Per Element=

Number of Diameter Loss Due EIRP

Elements of Element To Element (Including a11)

(Inches) Factor(dB) Losses) (dBW}

91 1.273 -3.065 55.191

127 1.077 -2.155 57.546

169 0.933 -1.601 59.338

217 0.824 -1.237 60.786

271 0.737 -0.986 62.002

331 0.667 -0.804 63.051

397 0.609 -0.668 63.976

.5 Watt

Magnification Factor= 4 Scan Loss=-2.139 dB Power Per Element= 1 Watt

Number of Diameter Loss Due EIRP

Elements of Element To Element (Including all)

• (Inches} Factor(dB) Losses) (dBW)

91 1.273 -3.065 58.201

127 1.077 -2.155 60.556

169 0.933 -1.601 62.349

217 0.824 -1.237 63.796

271 0.737 -0.986 65.012

331 0.667 -0.804 66.062

397 0.609 -0.668 66.986

Page z_-II
Table 4-1. EIRP for Array-Fed Dual Reflector Antenna (continued)

Frequency = 19.7 GHz Wavelength= .598

Diameter of Main Reflector = 56 Inches

Maximum Scan Angle= 3.5 Degrees

Inches

Focal Length = 56 Inches

Magnification Factor= 5

Number of

Elements

91

127

169

217

271

331

397

Scan Loss=-2.731 dB Power Per Element= .5 Watt

Diameter Loss Due EIRP

of Element To Element (Including all)

(Inches) Factor(dB) Losses) (dBW)

1.018 -3.065 54.600

0.862 -2.155 56.954

0.747 -1.601 58.747

0.659 -1.237 60.195

0.589 -0.986 61.411

0.533 -0.804 62.460

0.487 -0.668 63.385

Magnification Factor= 5 Scan Loss=-2.731 dB Power Per Element= 1 Watt

Number of Diameter Loss Due EIRP

Elements of Element To Element (Including all)

(Inches) Factor(dB) Losses) (dBW}

91 1.018 -3.065 57.610

127 0.862 -2.155 59.965

169 0.747 -1.601 61.757

217 0.659 -1.237 63.205

271 0.589 -0.986 64.421

331 0.533 -0.804 65.470

397 0.487 -0.668 66.395

Page 4-12

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characteristics of the former to be discussed in greater detail in the next

section, where the active aperture antenna de.sign and its performance

characteristics are addressed.

Dual Polarization. Two data streams operating at the same carrier

frequency can be multiplexed on the same dual-polarized feed horn The

circuit shown in Figure 4-2c schematically represents the use of

circularly polarized horns in the feed array. Qne data stream excites the

left-hand circularly polarized (LHCP) port of the horns; the other data

stream excites the right-hand circularly polarized (RHCP) port. Both data

streams modulate the same carrier frequency fl and use a different set of

phase shifters. Therefore, two independent beams are produced, operating

at the same frequency and each carrying its own data stream.

The feed horns used in a planar array inherently radiate a wave whose

polarization is very well-defined. The dual-reflector system tends to

degrade the polarization purity of the feed array. The degree of

degradation depends on f/D, surface accuracy, and beam scan angle. The

dual-reflector system may increase the amount of energy radiated in the

cross-polarized fields from less than -25 dB (with respect to the

copolarized maximum) at the feed array to as much as -15 dB in the

radiated beam. Generally, the cross-polarized energy in the main beam

(i.e., within the 4.3-dB beamwidth) will be less than -20 dB with respect

to the copolarized energy.

Page 4- 13
4.2. Active-Aperture Phased Array

An active-aperture phased array consists of a contiguous array of

antenna modules, each containing a radiating element excited by an SSPA

Each SSPA is driven by a modulator/driver through a diode phase shifter;

the latter is part of the antenna module. The phase of the signal fed to

each SSPA is adjusted by the phase shifter so that the radiated signals add

coherently to form a beam pointed in the desired direction. The diode

phase shifters can be programed to insert any phase shift, modulo 2_, in

less than a 0.1 I_sec. The active-aperture array is, except for overall size,

similar to the feed array described in the previous section.

Antenna Dimensions and Number of Elements. Consider a planar array

of horn antennas. Since the antenna beam has a circular cross section, the

array configuration should also tend to be circular. The elements of the

array are identical. Each element has a radiation pattern with an

approximately circular cross section and a half-power beamwidth, e3,

equal to or greater than the largest angle subtended by CONUS when viewed

from geosynchronous altitude. The elements are arranged on an equilateral

triangular, or hexagonal, grid and spaced to preclude the occurrence of

grating lobes pointing toward CONUS. Each horn has a circular aperture,

do, equal to the spacing between horns.

The approximate gain of the horn is given by (4-9) with M=1 The

maximum element gain at the edge of CONUS (Q = 3.5") is plotted in Figure

Page 4- 14

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4-3. This gain reaches a maximum value of 28.4 dBi for do/_. = 12. This

maximum is 4.3 dB below the element gain at the center of CONUS. For

do/_ > 12, element gain decreases. The ratio do/_ must be greater than 0.5

to support propagation. Therefore, 0.5 < do/_ < 12 and the corresponding

minimum element gain over CONUS varies from 10 to about 29 dBi.

The minimum gain of the array, Ge, occurs at the edge of a beam

coverage area and is given by (4-11) with M = 1, i.e.,

Ge(_)= 1.49N(lldo/_.)2[J1(u)lu]2 (4-16)

Equation 4-16 assumes that the antenna aperture is chosen so that the

4.3-dB beamwidth is equal to the angular diameter of the desired beam

coverage area. The 4.3-dB beamwidth and a major diagonal D of the

hexagonal array are related by

DI_.= 701e4.3 (4-17)

To find the value of N required to produce a given value of Ge in (4-16), it

is necessary to express do in terms of N through use of (4-13), (4-14), and

(4-17).

The EIRP of the active aperture array is given by

EIRP = 1.49PeN2(_do/_.)2[Jl(U)/U] 2 (4- 18])

Page 4-15
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Using (4-18), EIRP was calculated for different values of N, Pe, and D,

with maximum scan angle _ = 3.5". The results are shown in Table 4-2.

Note that the desired 0.9-deg beam coverage (i.e., 4.3-dB beamwidth) is

obtained with D --"46.4 in. instead of 56 in. as required by the array-fed

dual-reflector antenna. The difference in aperture size results principally

from the scan loss introduced by the dual-reflector system. The results

shown in Table 4-2 are plotted in Figure 4-4.

Radiation Patterns and Grating Lobes. The hexagonal active-aperture

array with 217 elements is shown in Figure 4-5. The diagram shows the

relative positions of the element centers; the element diameter is the

largest value permitted by the element spacing. Radiation patterns in the

plane of a major diagonal and in a plane bisecting an opposite pair of sides

can be calculated by reference to a linear array representing the hexagonal

array. In the first case, the off-diagonal elements in a direction normal to

the plane of the diagonal can be represented by a single-element radiator

with the appropriate excitation, as shown in Figure 4-5. The hexagonal

array can be represented by a different linear array to compute the

radiation pattern in a plane perpendicular to a pair of opposite sides.

An expression for the pattern E(8) in these planes is the product of

two terms, an element factor Geh(Q) and an array factor A(8). Exciting all

elements in the array with equal amplitude and a linear phase gradient to

scan the beam to the angle Ql, the radiation pattern in the diagonal plane

Page 4- 17
Table 4-2. EIRP for Active-Aperture Phased Array

Frequency = 19.7 GHz Wavelength= .598 Inches

Maximum Scan Angle= 3.5 Degrees

Array D_ameter = 58 inches Power per Element= .5 Watt

********************** Beamwidth= .72 ***********************

Number of Diameter Loss Due EIRP

Element_ of Element To Element (Including all)

(Inches) Factor(dB) Losses) (dBW)

91 5.273 -3.304 57.396

127 4.462 -2.319 59.825

169 .3.867 -1.721 61.662

217 3.412 -1.330 63.137

271 3.053 -1.059 64.372

331 2.762 -0.863 65.435

397 2.522 -0.718 66.370

Array Diameter = 58 inches Power per Element= 1 Watt

********************** Beamwldth= .72 ***********************

Number of Diameter Loss Due EIRP

Elements of Element To Element (Including all)

(Inches) Factor(dB) Losses) (dBW)

91 5.273 -3.304 60.406

127 4.462 -2.319 62.835

169 3.867 -1.721 64.672

217 3.412 -1.330 66.148

271 3.053 -1.059 67.383

331 2.762 -0.863 68.446

397 2.522 -0.718 69.380

Array Diameter = 52.2 inches Power per Element= .5 Watt

********************** Beamwidth= .8 ***********************

Number of Diameter Loss Due EIRP

Elements of Element To Element (Including all)

(Inches} Factor(dB) Losses) (dBW}

91 4.745 -2.640 57.144

127 4.015 -1.861 59.368

169 3.480 -1.385 61.083

217 3.071 -i.072 62.480

271 2.747 -0.854 63.662

331 2.486 -0.697 64.687

397 2.270 -0.580 65.593

Array Diameter = 52.2 inches Power per Element= 1 Watt

*********************** Beamwidth= .8 ***********************

Number of Diameter Loss Due EIRP

Elements of Element To Element (Including all)

(Inches) Factor(dB) Losses) (dBW)

91 4.745 -2.640 60.154

127 4.015 -1.861 62.378

169 3.480 -1.385 64.093

217 3.071 -1.072 65.491

271 2.747 -0.854 66.672

331 2.486 -0.697 67.697

397 2.270 -0.580 68.603

Page 4-I_

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Table 4-2. EIRP for Active-Aperture Phased Array (continued)

Frequency = 19.7 GHz Wavelength= .598 Inches

Maximum Scan Angle= 3.5 Degrees

Array Diameter = 46.4 inches Power per Element= .5 Watt

********************** Beamwidth= .9 ***********************

Number of Diameter Loss Due EIRP

Elements of Element To Element {Including all)

(Inches) Factor(dB} Losses) (dBW)

91 4.218 -2.062 56.699

127 3.569 -1.459 58.747

169 3.093 -1.088 60.357

217 2.729 -0.843 61.686

271 2.442 -0.673 62.820

331 2.210 -0.549 63.811

397 2.017 -0.457 64.693

Array Diameter = 46.4 inches Power per Element= 1 Watt

********************** Beamwldth= .9 ***********************

Number of Diameter Loss Due EIRP

Elements of Element To Element (Including all)

(Inches) Factor(dB) Losses) (dBW)

91 4.218 -2.062 59.710

127 3.569 -1.459 61.757

169 3.093 -1.088 63.367

217 2.729 -0.843 64.696

271 2.442 -0.673 65.831

331 2.210 -0.549 66.822

397 2.017 -0.457 67.703

Array Diameter = 40.6 inches

********************** Beamwidth= 1.03

Number of

Elements

Diameter

of Element

(Inches )

91 3.691

127 3.123

169 2.707

217 2.388

271 2.137

331 1.933

397 1.765

Power per Element= .5 Watt

***********************

Loss Due EIRP

To Element [Including all)

Factor(dB) Losses) (dBW)

-1.563 56.038

-1.109 57.937

-0.829 59.456

-0.643 • 60.726

-0.513 61.820

-0.419 62.781

-0.349 63.641

Array Diameter = 40.6 inches

********************** Beamwidth= 1.03

Number of

Elements

91

127

169

217

271

331

397

Power per Element= 1 Watt

***********************

Diameter Loss Due EIRP

of Element To Element (Including all)

(Inches) Factor[dB) Losses) [*BW)

3.691 -1.563 59.049

3.123 -1.109 60.947

2.707 -0.829 62.466

2.388 -0.643 63.736

2.137 -0.513 64.830

1.933 -0.419 65.792

1.765 -0.349 66.651

Page _-19
Table 4-2. EIRP for Active-Aperture Phased Array (continued)

Frequency = 19.7 GHz Wavelength= .598 Inches

Maximum Scan Angle= 3.5 Degrees

Array Diameter = 34.8 inches

********************** Beamwidth = 1.2

Number of

Elements

91

127

169

217

271

331

397

Power per Element= .5 Watt

***********************

Diameter Loss Due EIRP

of Element To Element (Including all)

(Inches) Factor(dB) Losses) (dSW)

3.164 -1.139 55.124

2.677 -0.810 56.897

2.320 -0.606 58.339

2.047 -0.471 59.559

1.832 -0.376 60.618

1.657 -0.307 61.554

1.513 -0.256 62.395

Array Diameter = 34.8 inches Power per Element= 1 Watt

********************** Beamwidth= 1.2 ***********************

Number of Diameter Loss Due EIRP

Elements of Element To Element (Including all)

(Inches) Factor(dB} Losses) (dBW)

91 3.164 -1.139 58.134

127 2.677 -0.810 59.907

169 2.320 -0.606 61.350

217 2.047 -0.471 62.570

271 1.832 -0.376 63.628

331 1.657 -0.307 64.565

397 1.513 -0.256 65.405

Array Diameter = 29 inches Power per Element = .5 Watt

********************** Beamwidth= 1.44 ***********************

Number of Diameter Loss Due EIRP

Elements of Element To Element (Including all)

(Inches) Factor(dB) Losses) (dBW)

91 2.636 -0.786 53.894

127 2.231 -0.560 55.563

169 1.933 -0.419 56.943

217 1.706 -0.326 58.120

271 1.526 -0.261 59.150

331 1.381 -0.213 60.065

397 1.261 -0.178 60.890

Array Diameter = 29 inches

********************** Beamwidth= 1.44

Number of

Elements

91

127

169

217

271

331

397

Power per Element= 1 Watt

***********************

Diameter Loss Due EIRP

of Element To Element (Including all)

(Inches) Factor(dB) Losses) (dBW)

2.636 -0.786 56.904

2.231 -0.560 58.574

1.933 -0.419 59.953

1.706 -0.326 61.131

1.526 -0.261 62.160

1.381 -0.213 63.075

1.261 -0.178 63.900

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is given by

Ed(e) : Geh(e)Ad(e) (4-19)

where Geh(e) is given by (4-9) with M -- 1 and _ = e, and

9

Ad(e) =9+_ aicos. __[ill(do/_.)(sine-sinel _ )]

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+ _-(18-i)cos [il_(do/1)(sin e-sin el)]

i=l

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where

ai = 8, i odd

ai = 9, i even

Note that a grating lobe occurs at eg when

Tl(dol_)(sineg-sin e I) = _+211 (4-21)

Solving for eg gives

sin eg = +_(2_./do) + sin e 1 (4-22)

The minus sign in (4-22) will be chosen, and e 1 will be assumed positive.

This combination, which places the grating lobe closest to the region of

interest (i.e., CONUS), yields

Page 4_23
sin eg = -[(2_/d o) - sin el] (4-23)

For the 217-elementactive-aperture array, do/)L= 4.66. Withe I = 3.5°,

eg = -21.7 °. The element pattern [see (4-9) with M=I] suppresses the

grating lobe 18 dB below the main-beam peak.

In the plane perpendicular to opposite sides of the hexagon,

E1(e) = Geh(e )Ap(e) (4-24)

where

8

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i=l

(4-25)

Note that a grating lobe occurs when

T_43(do/_,)(sin eg-sin 01 ) = +2_ (4-26)

Again choosing the minus sign,

sin eg = -[(2_L/_f3do)-sin t] 1] (4-27)

If the major diagonal is oriented to coincide with the maximum dimension

of CONUS, the maximum scan angle 81 in this case wit! be somewhat

smaller than 3.5", say 5.0 °. The corresponding value of eg is -tI.3" The

grating lobe at this location is suppressed 13.8 dB.

Page 4- 24

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4.3. MBA With Beam Switching Matrix

Several traveling wave tube amplifiers (TWTA) operating at 20 GHz

have been developed for communication satellite systems. Of prime

interest are a series that have been space qualified, have 25-30W average

power output, and operate at approximately 33-percent efficiency.

Manufacturers of these TWTAs claim that 50-60W devices are within the

state-of-the-art. Use of these tubes in redundant systems, withswitched

beam antennas, has forced development of ferrite latching isolator

switches that have less than 0.I dB insertion loss and handle up to l OOW

average power. Existence of these devices stimulates consideration of the

transmit antenna subsystem schematically shown in. Figure 4-6.

Signals from the payload signal router are fed to four modulators

which are connected to any four of eight TWTAs. A redundancy switch

connects the selected four TWTAs to four MBAs. Each MBA has a beam

switching network that connects a single input port to any beam port. The

latter ports are connected to feed horns, each of which produces a beam

pointing to a different 0.9-deg spot within CONUS. Beams radiated by the

four MBAs provide contiguous coverage over a 7 ° x 3" field of view that

spans CONUS. Dual polarization characteristics of the horns and operation

at either of two frequency bands permit simultaneous excitation of four

beams on a noninterfering basis.

Antenna Configuration. The beams from the MBA provide coverage

over a 0.9-deg-diameter area, with the coverage areas forming the

triangular lattice structure illustrated in Figure 4-7. Minimum gain over

Page 4-25
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Page 4-27
the coverage area occurs at points equidistant from three adjacent beam

peaks. The beam-to-beam spacing is J3 x 0.45" = 0.78". A total of about

60 beams is required to cover the 7" x 3" area spanned by CONUS. A

representative 42-beam configuration is shown in Figure 4-8.

To maximize the gain of the MBA and minimize feed spillover losses,

four MBAs producing an interlaced set of beams are used. Each MBA

produces beams arranged in a hexagonal lattice, but with beam-to-beam

spacing equal to 2 x 0.78" = !.56". This permits the feed horns for each

MBA to be spaced farther apart then would be the case if a single MBA

were used. The greater spacing between feed horns permits use of

larger-aperture feed horns, thereby minimizing spillover losses. This

results in approximately 3-dB greater gain than could be obtained with a

single MBA.

The gain of an MBA beam can be expressed as

Gm = Tl(_O/_)218(J2(u)/u2)]2 (4-28]

where D is the MBA lens diameter, 11is approximately 0.6, and

u = (11D/_,)sine (4-29)

The angle 8 is measured from the beam peak to the observation point.

The lens diameter, D, is chosen to maximize the gain at the edge of

Page 4-28
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Page #-29
the beam coverage area (i.e., 0 = 0.45"). As with the previously discussed

antennas, the 4.3-dB beamwidth is 0.9". The lens diameter is then

determined by taking the ratio of the gain at the beam peak to the gain at e

= 0.45"_

10 log [Gm(O" )/Gin(.45" )] = 10 log [8(J2(u.45. )/u2.45, )]-2

= 4.3 dB (4-30)

where u.45. is the value of u at e = 0.45". Solving this equation for u.45,

results in

u.45. -- (_D/_.)sin (.45") = 2.36 (4-31 )

Finally, D is determined to be 56.6. in. and the gain at the edge of the

coverage area is 43.0 dBi.

Assuming a loss of 1.5 dB in the RF switch network between the

transmitter and the MBA feed horns, a transmitter output power of i OOW

will produce an EIRP _ 61.5 dB over a beam coverage area 0.9" in diameter.

Switch Network. The switching design considered for use both in the

MBA beam-selection circuitsand the TWTA 8x4 redundancy switch is a

ferrite circulatorswitch configuration. This type of switch has been

space-qualified for satelliteapplications in the 20-GHz downlink band.

These switches have demonstrated low insertion loss and the abilityto

handle relativelyhigh (~100W) levels of RF power. The switching speed

Page 4-30

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depends on the amplitude of the voltage pulse applied to the circulator's

ferrite-core drive wire within the switch. For a given ferrite switch,

doubling the applied voltage will approximately halve the switching time.

Other parameters governing switch speed are the coercive force,

hysteresis properties, and the mass of the ferrite used in the switch.

A switch operating in the 20-GHz band requires about 7.54 iljoule for

switching. To switch in I00 ns, a 75-volt drive pulse is required. The

switch driver that produces this pulse can be expected to operate at about

15-percent efficiency, so the switching energy required is about 50 Iljoule

per switch per switching event. Thus a single switch, switching at 20K

events/sec, requires about one watt of driver power.

The 8x4 redundancy switch shown in Figure 4-6 requires 52 ferrite

switches, with a maximum of 5 switches in series between any TWTA and

the input to any MBA beam-selection switch network. The beam selection

networks require a total of 38 switches, with a maximum of 4 switches in

series between the input and any MBA feed horn. With 0.I dB insertion loss

per switch, and assuming 0.5 dB for the net loss of the interconnecting

waveguide, the total switch network loss is 1.4 dB, as previously

suggested.

Although the total number of switches required is 94, it can be

asasumed that the switches making up the 8x4 redundancy switch are

rarely, if ever, switched. In addition, to switch beams in the

beam-selection networks, a maximum of four switches in any network

Page 4-31
need be switched. A four-beam system will therefore require that no more

than 16, and usually only about eight, switches be switched to shift all

four beams. Thus, the maximum beam switching power for shifting four

beams at a 20K/sec switching rate is about 16W.

4.4. Comparison of Antenna Configurations

To achieve a specified performance level, the active-aperture phased

array and the array-fed dual reflector antenna require a similar number of

radiating elements. The former has the advantage of not suffering scan

loss. It also has better polarization purity in a dual-polarized system.

However, the direct-radiating array has a linear dimension that is larger

than the array dimension in a dual-reflector system by nearly the

magnification factor, M. Typically, M=3. Thus, the dual-reflector system

has the advantage of more convenient array packaging and perhaps more

conventional deployment. The two systems have comparable weight and

power requirements.

The MBA, on the other hand, requires considerably less power than

either of the other two configurations, because of the higher efficiency of

TWTAs as compared with SSPAs. It is estimated that this power ratio is

in the range of 50 to 60 percent. Aside from the need to develop a I OOW

TWTA at 20 GHz, however, the MBA approach suffers from the vulnerability

of the system to TWTA or beam-switching network failures. While TWTA

redundancy can ameliorate this vulnerability, reliable data on TWTA

lifetime is difficult to obtain. By comparison, both phased-array

approaches can tolerate a number of SSPA failures before system

Page 4-:32

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performance is noticeably affected. This graceful degradation greatly

enchances the attractiveness of the phased array configurations.

The MBA configuration also suffers from an aprior/ partitioning of

geographic coverage, as illustrated by the feed-horn assignments in Figure

4-8. If the traffic should not divide evenly among the four composite

geographic areas, saturation of a single downlink beam would lead to

degraded service at a total throughpu t which is less than the nominal

satellite capacity.

Because of the large number of elements involved, realization of

either phased array configuration at reasonable cost depends on advances

in monolithic microwave integrated circuit (MMIC) technology. The

desirable degree of integration (e.g., combined vs. separate modules for

each phase-shifter/amplifier pair) remains to be determined.

Page 4-3:3
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5. BULK DEMODULATOR TECHNOLOGY

The bulk demodulator units provide FDM/PSK-to-TDM digital

conversion and measure both signal amplitude and phase on over iO0

individualtone slots. The signalphase is differentiallydetected to form

the user data stream, while the signalamplitude is used during acquisition

for timing alignment. Both analog (SAW) and digital (FFT) bulk

demodulator techniques are examined, together with digitaldata detection

and post processing requirements.

Uplink symbol transitionsforeach FDMIPSK channel are synchronized

to the onboard bulk demodulator timing. This allows a factor-of-3

reduction in the number of Fouriertransforms required and eliminates the

need for data interpolation. If the terminals were allowed to be

nonsynchronous with the demodulators, many of the symbol start/stop

times would overlap transform boundaries and cause intersymbol

interference,as well as a loss indetected energy due to a time mismatch

in the detection filter. This time synchronization requirement is much

less stringent than for TDMA systems such as ACTS, where the terminals

must meet 25-ns accuracy. The accuracy requirement for the present

system is about 1 ps.

Intermodulation and adjacent-channel components are also a concern

with the FDM/PSK multiple-carrier format. With the SAW demodulator,

sufficient dynamic range must be provided to avoid nonlinear mixing

effects. In the FFT, the A/D and arithmetic quantization must be fine

enough to ensure accurate reconstruction of the signal, while the range

Page 5- !
must be wide enough to ensure that arithmetic overflow does not occur.

Since the transmit levels of the terminals in any uplink spot beam will be

approximately the same, the dynamic range can be limited to the number of

terminals multiplied by the signal-to-noise ratio for a single terminal.

With 100 channels and a received Eb/No of 15 dB, the quantization range

must be at least 35 dB. This requirement is easily met with an 8- to

10-bit A/D converter.

5.1. SAW Demodulators

One form of FDM/PSK demodulation uses surface acoustic wave (SAW)

devices to implement an analog spectrum analyzer known as a chirp

Fourier transform. Use of SAW demodulation for spaceborne

communications was first investigated by MIT Lincoln Laboratory for the

FEP EHF package on FLTSATs 7 and 8. Development of the SAW demodulator

at TRW has further refined the technology to meet the additional

capability and reliability required for MILSTAR. This effort has

demonstrated the ability to manufacture large numbers of these SAW

devices, as well as mechanical mounting and digital temperature

compensation techniques for both normal and stressed spacecraft

environments. The SAW demodulator typically requires from one-third to

one-fourth the power required by conventional LSl-based digital FFT

processors.

The SAW demodulator examined in this study consists of three basic

components: the Fourier transformer, a digital section, and a common LO

section (Figure 5-I). In this particular implementation, there are eight

Fourier transformer and eight digital sections for each common LO section.

Page 5- 2

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Page 5-3
Each demodulator path is responsible for demodulating 100 uplink user

channels, while the common L0 section supplies a chirp LO and digital

timing signals to each path. This LO sharing greatly reduces the overall

parts count and power, but still allows for graceful degradation in the

event of common LO failure. These components will be described in

greater detail in the following sections.

The Fourier transformer can be implemented in two generic forms:

convolve-multiply-convolve (CMC) and multiply-convolve-multiply (MCM).

These two transforms are mathematical duals of each other and, in theory,

produce the same results. The CMC transform, pictured in Figure 5-1,

provides additional input filtering and uses a single chirp LO generator,

thereby eliminating some of the temperature-controlled timing alignment

required in the MCM approach to hold phase stability.

SAW Fourier Transformer. A SAW Fourier transformer block diagram

is shown in Figure 5-2. The Fourier transform sections use devices known

as reflective array compressors (RAC), shown in Figure 5-3, to implement

dispersive time delay functions. The dispersive delays cause tow

frequencies in their bandwidths to be delayed longer than high-frequency

signals. Through use of RAC devices, linear FM chirps can be compressed

to a single pulse.

The Fourier transform section first passes the input signals through a

dispersive delay to offset their time positions and then mixes them with a

linear FM chirp from the common LO section. The mixing is done in a

bilinear mixer, which is implemented as a high-frequency analog

Page 5- 4

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multiplier to avoid intermodulation products. The result is a series of FM

chirps, one for each input signal, which cover the same sweep range but

overlap in time. Another RAC device, used as a chirp compressor, converts

these time-overlapped chirps into discrete time pulses, each

corresponding to a different input signal. The amplitude of each pulse

envelope and the phase of the carrier under the envelope depend on the

input signal amplitude and phase. The time position of each pulse is

proportional to the input signal frequency. Figure 5-4 shows the accuracy

of the reconstructed signal phase.

The output pulses from the Fourier transform section are routed to

the digital .section through a SAW bandpass windowing filter. Since the

SAW demodulator is a true Fourier transformer, input frequency is

translated to output time, and input time is translated to output

frequency. In particular, input-time offsets due to terminal timing

misalignment result in changes in carrier frequency under the pulse

envelope. Providing a filter characteristic in the frequency domain at the

output of the SAW Fourier transformer is equivalent, therefore, to time

windowing at the demodulator input. Windowing is desirable for reduction

of sidelobes caused by close adjacent-channel spacing.

The windowed pulses are passed through a limiter and a quadrature

phase detector. This removes any amplitude variations and produces

inphase and quadrature phase components which can be sampled and

processed by the subsequent digital circuitry. In addition, a linear power

detector is used to extract a measure of signal power for use in terminal

acquisition and temperature compensation.

Page 5- 7
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SAW Common LO. The common LO section provides the linear FM chirp

signal to each of the Fourier transform sections. The chirps are generated

by RAC devices which are sourced by an RF pulse generator. The pulses,

which consist of a few cycles of a high frequency LO (typically 200 MHz),

can be thought of as wideband (sin x)/x shaped bursts in the frequency

domain, with the sinc center at the LO frequency. The RAC devices delay

the low-frequency components of this signal by a greater amount than the

high-frequency components. This results in a chirp signal that shifts from

high to low frequency for each pulse input. The time positions of the input

signals at the Fourier transform mixer requires the generation of chirps at

a rate of one per input symbol and with a duration of almost two symbols.

This means that there are, in general, two different chirps simultaneously

present at the output of the chirp generator. The existence of two chirps

in the mixer LO port is the driving factor in the selection of a bilinear

mixer. Use of a conventional double-balanced RF mixer would result in

intolerably large intermodulation products.

Also located in the common LO section are the calibration tones and

the temperature compensation loop controller, which are used in

correcting demodulator temperature variations. The controller can be

shared among the eight Fourier transform sections and is described

further in the digital section.

Digital Section. The digital section input consists of a phase

quantizer, which performs quantizations based on a ratiometric

comparison of the I and Q inputs, and a sample/hold amplifier followed by

a low-speed, high-accuracy A/D converter, which can quantize the power

Page 5- 9
detector output for selected channels. The clock signals

quantizers are generated by a digital temperature tracking loop.

for the

Temperature compensation is necessary in SAWdemodulators because

the surface wave propagation velocity in the RAC devices changes as a

function of temperature. This causes the output pulses to vary position in

an accordian-like fashion, getting closer together or farther apart.

Careful design of the SAW demodulator transform relationships ensures

that this variation never causes one transformed time output to overlap

another.

Accurate sampling of a particular" set of frequencies, however,

requires a tracking loop that follows any changes in output time position

and corrects the quantizer sampling frequency and phase accordingly. The

tracking loop required in this application is quite similar to the loop

developed for MILSTAR. Calibration tones are injected by the common LO

section into eachFourier transformer, one at a time. The calibration tone

frequencies are well outside the portion of the input bandwidth used for

communications and are injected at the bilinear mixer. This produces an

output pulse that provides a reference time position for the band.

Switching between the lower and upper bandedges allows the temperature

compensation loop to track the time-offset and scale-factor variations

caused by temperature. A numerically controlled oscillator (NCO)in each

digital section provides a sampling which can be adjusted in both phase

and frequency to match that demodulator's characteristics.

The remainder of the digital section is devoted to data detection.

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QPSK data detection is performed by subtracting the phase measurements

of two adjacent symbols and comparing the result to one of four possible

90-degree phase shifts. The closest match is selected as the received

symbol and is passed to the downlink multiplexer. This differential

detection scheme provides carrier frecluency independence at the expense

of a doubling in BER. The resulting simplification in the terminal and

satellite hardware is considerable since no carrier phase coherence is

required.

SAW Bulk Demodulator Technology. SAW demodulators are currently

being manufactured for the MILSTAR program. These units have a slightly

different LO sparing arrangement, but use the same technologies.

Technologies developed at this point include: SAW RAC devices and

mounting techniques for 6-inch crystals, wideband bilinear mixers, SAW

window filters, log amplifiers, and a high-speed phase quantizer. The

temperature compensation hardware uses custom integrated circuits for

implementing the digital tracking loop and control functions.

Most of this technology is usable in its present form. However, new

RAC devices are required for a different application. In addition, a new

temperature control IC and microprocessor control algorithm are required.

Total power consumption for a current-generation unit is about 15W per

demodulation path, or 1500W for I00 demodulators. Significant

reductions in RF power required for the SAWs and the bilinear mixer are

not expected, but further reductions in sampler

power may reduce the power per demodulator to

demodulator is estimated to be 5 lb.

and control hardware

10W. The weight per

Page 5- 1I
5.2. FFT Demodulators

The FFT is an efficient implementation of the discrete Fourier

transform (DFT) in cases where a large number of orthogonally spaced

channels must be separated. The FFT is an all-digital implementation and

thus can be much lower in manufacturing cost than SAW techniques. In

addition, immunity to temperature .and aging effects make the FFT

desirable if low-power operation can be obtained. In this case, a

128-point FFT with 2:1 presumming is used to separate 100 channels at

approximately lO0-kHz spacing. The remaining 28 transform slots fall

into the roll-off of the input filter, which is used to prevent aliased

components of other channels from entering the desired l O0-channel

group.

An N-point FFT requires (N/2) log2N complex multiplications. For a

128-point FFT, this results in 448 complex multiplications. Since the

baud rate is roughly 50 kHz, with one FFT required per baud, the

computational requirement is 22.4 million complex operations per second

(22.4 MCOPS). Two different architectures can be used to perform these

operations, a recursive FFT and a pipelined FFT.

Recursive FFT. The recursive FFT, as its name implies, performs FFTs

using a single arithmetic element by recursively performing the algorithm

Since the FFT algorithm is regular in structure, it can be separated into

log 2 N passes of length N/2. An example of a chip set which currently

implements such a recursive FFT is the VHSIC Phase 1 FFT chip set, shown

in Figure 5-5 This TRW-developed chip set, which performs FFTs of

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variable length at rates up to 8.2 MCOPS, consists of a arithmetic unit, a

control unit, and several four-port memories. By the previously determined

sizing, three such sets would be required to perform the transform in the

required time. Actually, introduction of presumming and trigonometric

recombination functions (described later) make the need for a fourth chip

set probable.

Two members of the chip set have completed fabrication and initial

test. Current estimates are that the complete set will be available in

1987 and will consume approximately 2.5W of power. The total power

requirement for just the FFT portion is thus about 10W. Addition of the

A/D converter, buffering and control circuitry puts the total power at

approximately 12.5W.

Current-generation VHSIC Phase 1 FFT chips are radiation hardened

only to about 104 rad/Si, which is insufficient for most geosynchronous

requirements. Future generations of these chips, built in the radiation

hardened 2-1Lm CMOS process used for MILSTAR, should have radiation

hardening to 3x105 rad/Si and should also require less power The

projected total power requirement for a CMOS recursive FFT design is

about 9.5 W. The corresponding weight is estimated to be 3.5 lb.

Pipelined FFT. The pipelined FFT performs the same algorithm as the

recursive FFT, but it uses a cascade of high-speed stages to perform

successive passes of the algorithm in separate arithmetic units. This

procedure allows the input of the FFT to operate synchronously with

Page 5- 13
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the A/D sampling, thereby eliminating multirate clock difficulties often

encountered in recursive algorithms. In addition, the parallel (rather than

serial) processing eliminates the need to switch between processors at

each baud.

Figure 5-6 shows the architecture of a pipelined FFT developed at

TRW. The architecture is based on a complex multiplier, a radix-8 DFT

element, and a 64-sample intermediate memory. Each set of these

components can perform the equivalent of 15 MCOPS. Three sets are

required to perform the necessary calculations for a radix-128 FFT

(radix-8 x radix-8 x radix-2). These three sets can readily perform the

22.4 MCOPS previously established as the FFT requirement. These chips

require approximately 1.8W per set. They are currently manufactured in

the 2-1_m CMOS process, with some amount of additional processing needed

to produce radiation hardened devices. Total power for a pipeline design is

about 8.5W. Each unit weighs about 3.2 lb.

Real vs. Complex Sampling. Another issue for FFT-based demodulators

is the question of real versus complex sampling. In most textbooks, FFTs

are shown with complex input samples and complex results. This

arrangement eases understanding of the algorithms, but it often is not

practical to implement. For the present application, which involves

spectrum analysis, a high degree of phase and amplitude matching between

the quadrature input samples is required to eliminate introduction of

in-band al iasing.

As depicted in Figure 5-7, the complex-sample FFT relies on accurate

Page 5- 15
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cancellation of image frequency components. If the quadrature

downconversion process is unbalanced, these image components leak

through. This effect is especially degrading where the dynamic range

between users is high, since the image of a strong user can completely

mask a smaller user signal. A plot of image strength versus phase and

amplitude offset is shown in Figure 5-8. In this case, accuracies of better

than 0.5 dB and 2 degrees are required for adequate performance. These

tolerances, which are extremely difficult to maintain over temperature

and satellite aging, result in high design, component and assembly costs.

One technique for eliminating the effect of quadrature sampling

imbalance is to use real sampling. This requires that the input be

downconverted to a (low) frequency offset from baseband. The input is

then sampled by a single A/D converter operating at a rate equal to twice

the input bandwidth. Although the A/D speed is twice that of the complexsample case, only one converter is required. In addition, since there is

only one path, there is no concern for amplitude or phase matching with

this design.

Real sampling requires that input samples be fed only to the

real-component input of the FFT. With the same number of samples per

baud as complex sampling (half the number of samplers at twice the rate),

real sampling normally requires an FFT which is twice the size used for

complex sampling. In the present case, a 256-point FFT would be required.

This is feasible with both implementation approaches, but requires

additional hardware in the recursive design. The pipeline design shown

has sufficient speed capability to handle a 256-point FFT with the same

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hardware. The memory size must be increased, however, to accommodate

the larger number of transform points_

To avoid a factor-of-two growth in the recursive FFT architecture, a

technique called the trigonometric recombination algorithm was

developed. This algorithm allows every other sample of a real-sample

input to be placed in the I and Q inputs of the FFT. The in-phase and

quadrature components are transformed and, because of their orthogonal

input relationship, can be recovered at the output. An additional

algorithmic pass is required to combine image frequency components to

produce the desired result. Besides slowing the input rate seen by the FFT

by a factor of two, this procedure allows the" computations (and thus the

hardware) required to be reduced by a factor of two as well. The total

computational effort is approximately 14 percent greater than that

associated with a complex-input FFT.

Presumming. Since the FFT provides output samples at orthogonal

spacing (in this case about 50 kHz), while bi-orthogonal carrier spacing

(100 kHz) is used on the uplink, there are twice as many output values as

needed. This computational inefficiency can be reduced through a process

known as presumming. Figure 5-9 shows an 8-point FFT example.

If only the even-numbered channels are required at the output,

approximately one-half the computations can be eliminated by performing

only the additions (multiplication by WO) in the first FFT pass and then

implementing a 4-point FFT for the remaining passes. In the present case,

only 12B of 256 orthogonal positions are needed. Instead of implementing

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a 256-point transform, a 128-point transform with 2:1 presumming is

performed.

The presumming consists of storing the first-128 sample points

(complex) and then adding them one-by-one to the remaining 128 sample

points. This produces the 128 complex points which are the even

components of the first pass of a 256-point transform. A 128-point FFT

is then performed to yield the required results. This technique is

mathematically identical to the 256-point FFT and can be used to reduce

both the effective input sample rate and the FFT size required.

5.3. Conclusions

In terms of current technology, the SAW demodulator represents the

only technique which is both radiation hardened and currently undergoing

space qualification. Compared to current qualified discrete LSI

techniques, the SAW demodulator holds a favorable position for wideband,

low-power Fourier transforms. However, evolution of digital FFT

algorithms and LSI processes makes it quite likely that the eventual

choice for this application will be an FFT. Insensitivity to temperature

effects, constant high precision, lower power and reduced manufacturing

costs make FFTs an attractive alternative to SAW demodulators for future

applications.

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6. BASEBAND MULTIPLEXER TECHNOLOGY

The baseband multiplexer is responsible for assigning uplink

frequency channels to downlink time slots in a fashion which allows

programmable interconnection of terminals and efficient use of both

uplink spectrum and downlink data fields. A large number of serial

streams (representing the bulk demodulator outputs) must be multiplexed

into a small number of high-rate downlink streams with selectable paths

corresponding to particular downlink-beam dwells.

The requirements governing the multiplexer implementation trades

are summarized in Table 6-I. The number of multiplexer inputs

corresponds to the complement of I00 demodulators, each of IO-MHz

bandwidth. A nominal 100-kHz channel spacing has been selected for the

multiplexer analysis; consequently, there are I00 channels per

demodulator. The channel data rate of 413 x 64 = 85.3 kbps (which

includes the effect of rate-314 coding) has been rounded up to 100 kbps.

Consequently, the multiplexer inputdata rate is 10 Mbps per demodulator.

The corresponding multiplexer throughput is I Gbps. (Note that, when

rate-314 coding is appliedto the satellitethroughput of 790 Mbps, a coded

data rate of 960 Mbps results.)

It is assumed for descriptive purposes that the multiplexer must

simultaneously process one quadriphase symbol per channel for each of

10,000 channels. (Itwas pointed out in Section 2 that multiple symbols

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per channel must be processed jointly to limit the scan rate of the

downlink beams.) Six downlink beams are assumed for purposes of this

analysis. (The baseline system has four downlink beams.) There are

therefore 10,00016 = 1666 time slots per scan for each beam, under the

simplifying assumption that the two bits in each quadriphase symbol have

a common destination. The output data rate per beam has been rounded up

from 166 Mbps to 180 Mbps to account for the effects of overhead and

beam-switching intervals.

Three different multiplexing techniques were evaluated. Each was

selected based on its particular merits in one or more categories: size,

power, interconnection flexibility, and terminal constraints.

6.I. High-Speed SerialTime-Domain Multiplexers

A logicalfirststep in any multiplexer study isto consider the use of

standard, serial multiplexing techniques. Because of their serial nature,

these multiplexers are often small and easily understood. In this

application,however, serialmultiplexers are at a disadvantage. The high

rate of information flow, I Gbps, limits the abilityof serialmultiplexers

by requiring high-speed IC technologies which currently do not support a

high degree of integration. In addition,these multiplexers are often best

designed for single output applications where significant time

rearrangement is not necessary. In this application,not only must signals

be switched among several outputs,but they must also be rearranged in

time sequence to allow connection of any uplink frequency slot to any

downlink dwell.

Page 6- 3
The speed issue can be ameliorated by using a separate multiplexer

for each downlink. This lowers the maximum speed to about 170 Mbps and

eliminates many of the elastic buffering requirements which might appear

with several downlinks vying for the same multiplexer output. To

facilitate time reordering of symbols between bulk demodulator and

down]ink beam, each demodulator is provided with a 200-bit buffer

memory (one quadriphase symbol for each of 100 channels) which is

written in a random order and then read sequentially for downlink

transmission. The configuration of random write and sequential read

(rather than the reverse) is chosen to minimize speed requirements during

downlink readout, where successive reads may be spaced by as little as 6

ns. (Interleaving addresses on smaller memories could reduce the speed

requirements further, but at the expense of large numbers of extremely

small memories.)

With a single 200-bit buffer assigned to each demodulator, the

possibility exists that two or more downlinks may require symbols to be

read from a commonbuffer memory for transmission during the same time

slot. This memory contention problem can be avoided by providing a

separate buffer memory for each demodulator/downlink pair. The most

logical configuration would be a set of six addressable shift registers, as

shown in Figure 6-1. These could be integrated on the same chip and

fabricated in a high-speed bipolar process.

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Combining of demodulator outputs can be done in either a serial chip

interconnectionscheme, shown in Figure 6-2, or a parallel multiplexing

scheme, as shown in Figure 6-3. The serial scheme requires fewer

interconnections,but requires relay cross-strapping of the serialchain at

regular intervalsto avoid single-pointfailures.

6.2. Frequency-Domain Multiplexers

A simple technique for reducing on-board multiplexing requirements

isto arrange the uplink frequency slots in a manner that allows simple and

efficient direction to the downlink. Reassignment of multiplexed

interconnections is performed by reassignment of uplink frequencies and

downlink time slots.This results in minimum on-board processing, but it

puts some limitationson interconnectionflexibility.

Figure 6-4 shows an uplink frequency assignment plan for an

approximately 80-MHz segment (one-sixth) of the 500-MHz uplink band.

The plan is designed for a satellite system with six independent downlink

scanning beams (labeled A through F). This frequency plan is arranged so

that each sixth frequency slot is assigned to the same downlink beam.

This allows simple uplink spectrum reorganization by downlink beam. It

also helps reduce multiplexer speed requirements, since similarly lettered

bits from each demodulator can be passed in parallel to a common buffer

and converted to serial form for high-rate downlink transmission.

Furthermore, adjacent 80-MHz uplink segments have their A to F

downlink beam assignments rotated by one position, thereby ensuring that

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symbols transmitted simultaneously on the six beams are drawn from

different 80-MHz segments. A particular downlink beam, say A, is fed

data from the lowest 80-MHz segment (on both polarizations) during the

first uplink symbol interval. It is fed data from the second segment during

the next uplink symbol interval, and so on until all six segments have been

read. At this point, the downlink multiplexer returns to the first 80-MHz

segment and the sequence is repeated.

Approximately 16 demodulators are read in each position, eight from

each uplink polarization. As the A multiplexer moves to the demodulators

that correspond to the second 80-MHz segment, the B multiplexer moves to

the first 80-MHz segment to receive symbols from the B-designated

frequency slots. In this manner, a continuous flow of data is maintained

and the word transfer rate does not exceed I0 MHz, allowing CMOS

technology to be used for low power consumption.

The description thus far has focused on the association of uplink

frequency channels with particular downlink beams. The nature of the

traffic matrix supported by such a multiplexing scheme is also illustrated

in Figure 6-4. On the uplink, a specific set of demodulators supports each

regional beam. One-sixth of the composite capacity of these demodulators

is directed at each downlink. Within any frequency subset (corresponding,

for example, to letter A), the fractional number of channels allocated to

(say) New York-directed traffic must correspond to the percentage of time

the downlink beam dwells on New York. This is true for every uplink

region.

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Whether this constraint on the allocation of satellite capacity

significantly decreases the traffic throughput depends largely on the

number of uplink regions. The effect on throughput will be relatively

small ifthe number of uplinkregions is small. (Note that there are eight

uplink regions in the baseline system.) However, ifa single metropolitan

area (such as New York) dominates a region,the percentage of trafficfrom

that region to the metropolitan area may be significantlyless than the

percentage of trafficfrom other regions to that metropolitan area. In this

case, the corresponding satellitecapacity will be unclerutilized.

A further shortcoming of thismultiplexing scheme is that continuous

transmission on some channels may have to be interrupted for short

intervals during reallocation of satellitecapacity. This is a drawback

when compared to a truly random assignment multiplexer,but it is not a

severe constraint if only short-duration data/voice connections are

required.

6.3. Memory-Based Multiplexers

Memory-based multiplexing techniques are gaining increasing favor

because of the tremendous flexibility that they offer (random

interconnect).Memory technology is also a rapidly developing area, with

increased size,faster speed, and lower-power units being introduced at a

quickening pace. Current concerns with spaceborne memory devices center

on radiation hardness and single event upset (SEU). Many current CMOS

memory devices are showing satisfactoryhardness (IE5 radlSi) for space

environments, but they are limited to about lO0-ns access times.

Page 6- I1
SEU-induced bit flips do not represent a significant problem for this

application, because they show up as random bit errors which are

corrected by the end-to-end coding. SEU latchup remains a critical

concern in device selection. Current generation, hardened memories are

designed to be latchup-immune through use of dielectric isolation,

epitaxial processes, or similar techniques.

Memory-based multiplexers range in size and complexity from simple

address-pointer techniques to time-space-time techniques similar to

those found in telephone switching systems. The address-pointer

technique is shown in Figure 6-5. This is a modification of the serial

multiplexing technique. However, in this case each demodulator has a

memory which can hold up to 2 x 104 bits of data. This memory capacity

represents one data symbol for each downlink-beam/time-slot

transmission that takes place during a particular uplink symbol interval.

Each demodulator also has a 1O0 x 14 bit RAM which acts as a map to place

each bit from the bulk demodulator in a particular memory position.

Broadcast capability can be implemented by placing individual uplink bits

into more than one buffer location. The buffer RAMs are then read in

60-bit-wide words which are wire-ORed together in a trellis network to

derive six sets of l O-bit words at a 17-MHz rate. These are then

serialized for transmission at the downlink rate. The wire-OR function

serves to passively multiplex the uplink bits together and can be designed

to be single-point error immune. Active selection of necessary bits from

the bulk demodulator memories was also considered, but was ruled out

based on the high speeds required in the memory read.

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An alternate, and very useful, technique involves serial read-out

memories which can be read at rates much higher than the write speed (by

performing internal parallel-to-serial conversion). The techniques for

producing such memories have been developed commercially to satisfy the

need for high-rate raster-scanned computer displays and could be applied

to the multiplexer example directly through incorporation of

bit-addressable inputs and fabrication in radiation-hardened processes

This greatly reduces the number of external OR operations required and

makes serial "daisy-chaining" of chips possible, if reliability concerns can

be alleviated through cross-connect switches. A diagram is shown in

Figure 6-6.

These memory-based schemes have drawbacks in terms of the

efficiency of on-board memory utilization, which is approximately one

percent. The total memory required is about 2 Mbits. However, the low

cost and versatility of memory-based structures makes such techniques

attractive.

Several conclusions can be drawn from examination of the various

multiplexing techniques. If on-board simplicity is most desired, the

frequency-based assignment technique involves minimum on-board

processing and no on-board reconfiguration other than dwell times. If

random interconnect, broadcast capability, and lack of outages are most

important, the memory-based scheme would seem to have considerable

merit. Since the latter properties are most desirable in a thin-route

network, the memory-based structure is the preferred configuration

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7. TRAN5MITTER AND RECEIVER TECHNOLOGY

Design of the satellite uplink and downlink are heavily dependent on

achievable performance in the areas of high-power amplifiers and

low-noise receivers, at both 30 and 20 GHz. This section provides an

assessment of current technology in these areas, as well as projections of

performance in the 1990 and 1995 time periods. Only solid-state devices

have been considered for the present application.

In assessing 20-GHz power amplifier performance, the quantities

normally of interest, power rating and DC/RF conversion efficiency, are

discussed. However, it should be pointed out that in the baseline satellite

design, each transmit antenna is an array-fed reflector configuration

involving 331 elements. Each element has associated with it a final

amplifier that produces 300 mW of RF power. Therefore, 20-GHz

high-power devices are not especially important to the baseline design.

By contrast, high DC/RF conversion efficiency is most important.

Amplifying device power levels are shown in Figure 7-1 for both 20

and 30 GHz. Corresponding device efficiencies are shown in Figure 7-2. In

addition to indicating currently achieved performance, various vendors

provided performance projections for 1990 and 1995. Except where

indicated otherwise, all devices are GaAs FETs. In 1990, single-device

power levels of 4W can be expected at 20 GHz, and 1.5W to 3W at 30 GHz.

Corresponding collector (i.e., DC/RF) efficiencies are 35 percent at 20 GHz

and 25 percent at 30 GHz.

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IMPATT diodes are the only devices that today can provide 2W of

power at 30 GHz. With similar power levels anticipated for GaAs FETsby

1990 or shortly thereafter, the latter will become the technology of

choice for °low-power applications. FETs, being three-terminal devices,

are inherently more stable than IMPATT diodes. In addition, they are

capable of higher efficiency. Finally, they offer the potential for

monolithic manufacture, which can lead to greatly reduced cost.

The potential exists for substantial increases in device efficiency by

1995. HEMTand GaAs FET device efficiencies of 45 percent have been

projected by different vendors. Of the two technologies, GaAs FETsshould

have the greater power capabil ity.

Translation of the device efficiency of Figure 7-2 into amplifier

efficiency dependson the required amplifier gain. Two examples are given

in Figure 7-3, using the Raytheon 20-GHz device projections for t990. If

the 6-dB gain available from a single stage suffices, the DC/RF efficiency

is identical to that cited in Figure 7-2 at the device level• TIne

power-added efficiency is only 26.2 percent, however. In the second case,

a three-stage gain of 15 dB is required. Because of losses introduced in

the first two stages, the DC/RF efficiency is reduced to

However, the power-added efficiency of 24.0 percent is

below that of the first case.

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be rnultiplied by a factor of 1.25 to account for an 80-percent power

conversion effic iency.

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Vendor projections of LNA device performance are given in Figures

7-4 and 7-5 All devices are GaAs FETs, except where HEMT devices are

indicated. By 1990, noise figures on the order of 1.5 dB are widely

predicted at 30 GHz, with slightly lower figures at 20 GHz. Device gain

values of 10 dB should be achievable at both frequencies by 1990.

Longer-range HEMT performance projections by TRW are considerably

more optimistic than long-range vendor projections for GaAs FETs. By

1995, GalnAs HEMT noise figures as low as 0.5 dB are foreseen at both 20

and 30 GHz (Figure 7-6). The anticipated gain of these devices is

relatively low, however

The noise figure for a complete amplifier depends on the number of

stages that are combined and the per-stage gain. Based on achieved (1986)

TRW HEMT performance, a two-stage amplifier would have a noise figure

of 1.4 dB (Figure 7-7). Because of the relatively high per-stage gain of !2

dB, the overall noise figure is only 0.1 dB higher than that of a single

stage.

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8. SATELLITE WEIGHT AND COST

Prior to cost estimatioh, weight and power estimates are obtained

for each satellitesubsystem. For certain major payload elements (e.g.,

receiverldemultiplexer section),a unit weight and power breakdown is

provided. For the demodulators and the multiplexer, the chip complement

is examined to obtain a power, and thereby a weight, estimate.

8.I. Payload Weight

For purposes of weight and power estimation, the payload is

subdivided by major function as shown in Figure 8-I. Accordingly,

estimates are provided for: receiverldemultiplexersection,demodulators,

baseband processor (i.e.,multiplexer),modulator, and transmit arrays. A

weight and power summary for the payload is given in Table 8-I. Details

of these estimates are provided below.

ReceiverslDemultiplexers. The receiverldemultiplexer section of the

satellite repeater accepts user transmissions from each of the eight

CONUS regions,downconverts the received signals to baseband, and routes

the signals to the appropriate bulk demodulators. A functional block

diagram of this section of the repeater is shown in Figure 8-2. Only the

portion corresponding to a singlepolarizationis shown.

An individualdemodulator accepts a composite signal bandwidth of

I0 MHz. Each of the 50 demodulators associated with a particular

polarizationmust be connected tothe uplinkbeam to which its IO-MHz

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segment has been assigned. It must also be provided with a

downconversion chain that translates the IO-MHz segment to baseband

(i.e.,1-11 MHz). It is assumed that,of the fifty IO-MHz segments on each

polarization,32 are permanently assigned to particularuplink beams. For

simplicity, these 32 segments are shown in Figure 8-2 as distributed

uniformly among the four uplink beams and assigned to demodulators 1 to

32. The remaining 18 IO-MHz segments (i.e.,demodulators 33 to 50) can

be assigned to any of the four uplinkbeams by means of the 4 x 28 switch.

The connections are chosen to match the time-varying geographic traffic

divisionamong the four beams.

In addition to the first 50 demodulators, which are given fixed RF

frequency assignments, !0 spare demodulators are provided for each

polarization. The downconverters preceding these demodulators have

variable LO frequencies. The spare demodulators also serve to avoid

reduced channel utilization due to a division of the uplink-beam traffic

which does not correspond to an integer multiple of 10 MHz in each beam.

For example, to divide a given IO-MHz segment between two beams, each

beam output would be connected to a demodulator assigned to that I O-MHz

segment. One of these demodulators would be selected from the "spare"

complement. At the demodulator outputs, only those samples

corresponding to occupied channels would be fed to the downlink beams.

Weight and power estimates for the units in Figure 8-2 are given in

Table 8-2. The only nonredundant units are the 4-GHzlbaseband downconverters, inasmuch as spare units are explicitlyprovided. Units that

support half the demodulator complement or less have been made singly

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redundant. Those units supporting both polarizations (i.e.,the 6.5-GHz

frequency source and power conver[ers I and 3) have been made doubly

redundant.

In a number of cases, unit sizing is based on existing designs.

example, the 3014-GHz downconverter is similar to

ACTS downconverter. The power converters are

generation MILSTAR designs.

For

the corresponding

based on secondThe weight and power of the receiverldemultiplexer section are

dominated by the two frequency sources (one for each polarization).These

units contribute 41 percent of the total weight and 37 percent of the total

power. The technology involved in the frequency sources is not expected

to change radically in the future, so significant weight and power

reductions are not expected. In addition, the 120 4-GHzlbaseband

downconverters account for 26 percent of the required power. Significant

reduction inthis value is also notanticipated.

Demodulators. Demodulator weight and power estimates are provided

for a pipelined FFT configuration (see Section 5.2) using 1-11m CMOS VL51

technology. Sizing is based on 1991 technology estimates for L51 density

and power consumption.

The requirements for a 128-point FFT with real sampling, as

established in Section 5.2, are:

• 24.6-MHz sampling rate (real samples)

Page 8-7
• 128 points with windowing and trigonometric recombination

• lO-I_S computation time

These requirements translate into (NI2T)(Iog2N + 2) equivalent complex

operations per second (COPS), or 57.6 MCOPS per demodulator.

As a basis for estimate, two current FFT technologies were examined:

the TRW VHSIC I FFT chip set and a TRW pipe]ined FFT chip set. The VHSIC

chip set comprises a l-pro CMOS control chip, a 1.25-LLm CMOS four-port

memory, and a 1.25-_Lm CMOS arithmetic unit chip. This chip set is

capable of 8.192 MCOPS at a power dissipation of 2.2W. Improvements

expected in 1987-88 are expected to reduce this value to about 1.8W. The

equivalent power per MCOPS is 220 mY/.

The pipelined chip set uses 2-1_m CMOS technology and consists of

four chip types: a complex multiplier, an 8-point discrete Fourier

transform (DFT), a control-chip, and a 674-word memory. This chip set is

capable of 15 MCOPS and dissipates approximately 2.5W Future

conversion to 1-11m CMOS is expected to reduce power consumption to

1.5W, or 100 mW per MCOP. Future advances in areas such as submicron

wafer-scale integration (W51) are expected to reduce power consumption

to values below 1O0 mW per MCOPs in the ! 992 time frame.

Based on the power requirements of a pipelined chip set using l-pro

CMOS technology, the power needed for a 128-point FFT is 5.8W.

Additional input/output memory for the FFT requires about 500 roW, based

on 4K bits of storage at 25 MHz and a l-pro CMOS estimate _f 0.1 mWlbit

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for hardened static RAM. Among other demodulator elements, the 25-Msps

8-bit A/D converter currently requires 1.2W; however, improvements in

CMOS processing should reduce the power level to 600 mW. Postprocessing data detection, accomplished by a semicustom 1-11m CMOS

chip, consumes about 250 mW. Finally, control and logic overhead and

interfaces require about 1W. Thus, the total power needed for a single

demodulator is 8.15W.

A complement of 100 active demodulators would therefore require

815W of secondary DC power. Assuming 80-percent-efficient power

converters, the primary DC power required by the demodulators is 1020W.

The demodulator weight is based on a rule-of-thumb coefficient of

3.3 W/lb for CMOS satellite hardware. For purposes of weight estimation,

the spare demodulator units must be included; therefore, the appropriate

secondary power is 1.2 x 815 = 978W. The equivalent demodulator weight

is about 300 lb.

Multiplexer. The multiplexer is based on a three-level memory

multiplexing technique with an equivalent of three chips per demodulator.

These chips have high-speed interfaces with each other and with the

demodulators. In addition, bipolar, high-speed formatter chips are

required for each downlink.

Each memory chip consists of a number of serial inputs and outputs,

together with a multiport internal-registered data memory and a separate

control table memory. These chips are constructed in l-I_m CMOS and

Page 8- 9
consume 350 mW. The output formatters require 3 to 4 chips per

downlink, at approximately IW per chip. Onboard computer control is

expected to require lOW. With I00 active demodulators and four

downlinks, the multiplexer requires about 130W of secondary DC power.

The power requirement on the primary side of the power converters is

165W.

The multiplexer weight is computed from the secondary DC power

that would be required for 120 active demodulators, which is 152W. At

3.3 Wllb, this power level translates into 46 lb. Because of the bipolar

components, which have a higher power/unit-weight ratio than CMOS, the

multiplexer weight may be taken as 40 lb.

Modulators..The modulator section of the repeater is shown in Figure

8-3. The four BPSK modulators have as inputs the appropriate data

streams for the four downlink beams and a 5-GHz tone. The latter

undergoes a x4 multiplication in the modulator to provide a 20-GHz

carrier. The 20-GHz solid-state amplifier raises the carrier level to the

required input for the initialamplifier in the HPA chain. The latter

successively divides and amplifies the modulated carrier until the

required number of signals (331 in this case) are provided to the radiating

elements at theappropriate power level(0.3W). The modulator weight and

power estimate is detailedby unit inTable 8-3.

Transmit Arrays. There are two identical transmit arrays, each

supporting two of the four downlink beams. The elements of each array,

together with weight and power estimates, are itemized in Table 8-4. The

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baseline array design has 331 radiating elements. Since a separate set of

phase shifters is needed for each beam, there are 662 phase shifters in

each array. There is a like number of final amplifiers; however, an

additional iO0 or so amplifiers are needed for the divide-and-amplify HPA

chain.

The DC power requirements for the HPAs are based on an assumed

25-percent efficiency and a radiated power per element of 0.3W. Phase

shifters for a phased array at 20 GHz do not presently exist; consequently,

estimates of required power are subject to sizable errors. Based on

discussions with workers in this field, it is felt that the estimate of

0.25W per phase shifter used in Table 8-4 is realistic, and may in fact

.prove conservative. The phase-shifter controller power estimate is based

on use of PROM look-up tables for each element. This approach is made

feasible by a time-tagged power strobe technique, in which each PROM is

powered only when it is being read. This process requires less than I I_s.

8.2. Satellite Weight

As shown in Table 8-1. the baseline payload configuration weighs 975

lb and consumes 4955 of power. The satellite power requirements are

detailed in Table 8-5. It is assumed that 25 percent of the full

communications capability is maintained during eclipse. This choice

results from the tradeoff of satellite weight vs. eclipse capability

displayed in Table 8-6.

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Eclipse Capability(%) Satellite Weight(lb)

10 5310

25 56O3

5O 6O93

100 7073

The power consumed by the payload and supporting subsystems, as

seen at the interface with the electrical power subsystem (EPS), is

6080W. When power distribution losses and the need to recharge the

batteries during normal operation are taken into account, the power

requirement increases to 6850W. The corresponding EPS weight is 1800

lb. In computing the battery charging requirements, an eclipse period of

1.2 hrs is assumed. The battery sizing is based on (I) specific weight of

140 ib/kW for NIH 2 batteries, and (2) 75 percent depth of discharge.

A satellite weight budget is given in Table 8-7. The payload and EPS

weights have already been accounted for. The thermal control subsystem

weight is explained in Table 8-8. All heat-generating units other than the

transmit arrays are presumed to require cold plates and a heat transport

mechanism, in addition to radiators. Only radiators are required by the

phased arrays. The 2235W dissipated by the pair of phased arrays

accounts for the difference in load presented to the thermal subsystem

elements.

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Table 8-8. Thermal Control Subsystem Weight

Specific

Item Weight (IblkW) Net Load (W) Weight (Ib)

Cold Plates 38 3095 118

Heat Transport 19 3095 59

Radiators 75 5330 400

Total 577

The propulsion subsystem weight is based on the need for 50 Ib of

propellant per year for orbit maintenance over a period of 10 years. In

addition,150 Ib of propellant is provided for insertion/initialization,and

50 Ib for emergency disposal. The structural weight of the propulsion

subsystem istaken as II percent of the propellantweight.

Examination of Table 8-7 reveals that the satellite weight is largely

attributable to the payload power requirements. It is tempting, therefore,

to reduce the power requirements through an increase in the size of the

transmit arrays (i.e., an increase in the number of elements with the

inter-element spacing held fixed). For example, if the number of elements

is doubled, the RF power requirement is halved, and the DC power

consumed by the power amplifiers is also halved. However, the DC power

associated with the phase shifters is doubled, under the assumption that

each amplifier output has an independent phase setting. The net result is

that the DC power required by the pair of phased arrays decreases from

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2980W to 2230W. The combined array weight, however, increases from

300 lb to 540 lb. Insofar as total satellite weight is concerned, the

decreased power and the increased weight of the transmit arrays offset

one another; i.e., the two configurations have virtually identical weight.

It is possible,with double the number of radiatingelements, for each

phase shifter to serve a pair of elements, thereby holding constant the

number of phase shifters. (Gratinglobe and scan loss properties must be

re-examined, however.) This further reduces the DC power consumed by

the phased arrays from 2230W to 18 lOW. However, the satelliteweight is

only reduced from 5620 to 5460 lb..Moreover, the resulting satellite

configuration (as well as the configuration with twice the number of

phase shifters)has the disadvantage of significant added complexity in

the phased arrays. For these reasons, the satelliteconfigurationbased on

a pair of 331-element phased arrays is retained as the baseline design.

8.3. SatelliteCost

Engineering estimates of the satellite cost presented below are

exclusive of fee. Estimates of nonrecurring and first-unit costs are

provided. Three separate assembly phases are assumed: payload, bus, and

payload-to-bus. A satellitecost summary is shown in Table 8-9. (Listing

of the cost estimates to one decimal place is a computational convenience

and is not intended to imply knowledge to that accuracy.) The nonrecurring

or development cost is $I 10 million,while the first-unit cost is $I 13

million. All costs are in 1987 dollars.

Page 8- 19
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Cost estimates for the payload hardware elements are based on a

dollars/pound analogy with space-qualified electronics of known cost.

The payload cost is compiled in Table 8-I0 by enumerating, in addition to

the major hardware sections (e.g.,receiver/demultiplexer),the activities

required to delivera fullyassembled payload. For those hardware sections

involving a large number of similar units (e.g.,the bulk demodulator

section has 120 distinctdemodulators), the term "quantity"refers to the

number of hardware "slices" into which the section is divided. For

example, the bulk demodulator section comprises 15 slices, each

containing eight demodulators and weighing 20 lb. Each of these slices is

considered a unit for costing purposes. The firstfiveunits are assumed to

be prototypes, all of which are produced at identicalcost. A g5-percent

"learning"curve is assumed to apply to the remaining units.

The nonrecurring cost (NRC) iscomputed according to

NRC = 3(RC) + 0.25 (N-1)(RC)

when RC is the recurring cost and N is the number of units produced.

The spacecraft bus cost estimate is given in Table 8-11. Spacecraft

subsystem costs represent best estimates based on comparisons of cost

estimating relationships (CER) and dollars/pound analysis.

Page 8- 21
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9. USER TERMINAL DESIGN AND COST

Based on the system architecture described in Section 2, the user

terminals must satisfy the followingrequirements:

• Transmit a single64-kbps, QPSK carrier

• Frequency agilityover 500 MHz

• Transmit (on command) either right-hand or left-hand circular

polarization;receive on oppositesense polarization

• Transmitter power of 2W

• Figure-of-merit,G/T, equal to 18.3 dBIK

• Differentially detect BPSK data bursts on 240-Msps carrier;

received carriermay be ateither of two frequencies and on either

polarization

• Provide voice and/or data capability.

In addition to these technicalrequirements, the terminal design was

influenced by the following factors:

• Minimization of (recurring)cost

• Terminal production in largequantities(e.g.,10,000 units)

• Deployment to begin in early IggOs.

Accordingly, the following quidelinesand assumptions were observed

in the terminal design procedure:

• Maximum sharing of uplink/downlink resources

Page 9- 1
• Functional partitioning to minimize environmental control needs

• Minimum number of frequency translation stages

• Nobuilt-in test equipment or h_alth status reporting

• Maximum use of automated factory procedures, including

automated alignment and testing

• Temperature control optional

9 1. Terminal Description

The terminal block diagram, shown in Figure 9-1, is partitioned into

six subassemblies, which are designated AI through A6 The downlink

signal path is shown in bolder lines than the uplink path. Figure 9-2

depicts the terminal packaging and indicates where the subassemblies

reside

Subassembly A1, which is one of two indoor subassemblies, performs

the following functions:

• A/D multiplexing for voice digitizing and data routing, according

to the user-selected mode

Satisfaction of network requirements for coding, frame

formatting, and data transition and frame-time slot acquisition

under CPU control

• QPSK modulation at 70 MHz

• Provision of high-precision and low-phase-noise I O-MHz reference

for uplink and downlink frequency sources

Page 9- 2

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• UPLINKIF &LO REFERENCE

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• Provision of electrical-to-optical conversion for both the I O-MHz

reference and the 70-MHz QPSK signal.

Data transition times are controlled because it is necessary that

these transitions occur simultaneously in all channels accessing a common

satellite bulk demodulator. A frame structure is established so that

periodic orderwire bits arrive at the satellite simultaneously.

Indirect, rather than direct or RF, modulation was chosen because

direct modulation is relatively expensive and is attractive only when size

and weight are of paramount importance. A direct modulator would reside

in the outdoor package and would impose the need for strict temperature

control. Indirect modulation is performed indoors at a common IF

frequency.

Costs associated with high frequency stability dictate the use of a

single frequency reference, together with a distribution network to all

reference sources within the terminal. Reference oscillator costs are

very sensitive to temperature range requirements. To reduce this cost

driver, the frequency reference is placed in the indoor electronics.

Both copper and optical fiber cabling were considered for the

indoor/outdoor interface (Subassembly A3). The costs of the two

materials are now comparable. However, fiber has the following virtues,

which led to its selection:

• Signal sharing of a relatively small and flexible cable

• Immunity to electrical interference

Page 9- 5
Availability of cost-effective LEDsources to support the data rate

requirements

• Removalof indoor/outdoor distance limitations

• Rapidly dropping

•conversion costs.

electrical/optical and optical/electrical

The uplink IF signal is converted to S-band in Subassembly A4. The

final upconversion is accomplished using a harmonic mixer (n=2) and an

OEMS-band synthesizer and X6-multiplier second LO. The OEMsynthesizer

provides the required uplink frequency agility, while selection of a

harmonic mixer, with input at 13.8 GHz, eliminates the need to provide

costly gain at 26 GHz. Pre- and post-HPA filtering is included to meet FCC

spurious emissions requirements. The HPA is a 2W IMPATT amplifier with

on/off control via a power gate controlled by the indoor CPU (part of

Subassembly A2). The 30-GHz transmitter output from the A4

subassembly (contained in a waterproof housing behind the antenna) drives

the transmit feed via a circular polarizer, which is controllable to either

RHCP or LHCP.

The received signal is first passed through a low-pass filter, which

provides isolation between transmitter and receiver. The filter output is

fed to Subassembly AS, which is contained in a weatherproof housing

behind the antenna. The LNA and associated filters (for rejection of

out-of-band signals) exhibit a noise figure of 2.6 dB. Single conversion to

a 4-GHz IF is accomplished by a harmonic mixer (n=2) with an LO input

derived from a X4 multiplier fed from the uplink first LO.

The received carrier can be at either of two frequencies. Separate

Page g- 6

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DBPSK demodulators, with center frequencies of 3.88 and 4.02 GHz, are

provided for the two cases. Selection of the appropriate demodulator

output is done at baseband. A single demodulator with transfer switching

to accommodate the two frequencies could be used instead, at comparable

cost. However, hardwired, dedicated demodulators should prove more

reliable.

The received signal power variation will be no greater than 10 dB.

This variation is small enough to be removed at baseband, thus precluding

need for a costly voltage-controlled attenuator at 4 GHz.

Clock recovery is a key issue. Because symbol transitions from burst

to burst are spaced by multiples of the baud length, clock recovery during

startup can take place over many bursts, with a "flywheeling" technique

used to recognize symbol transitions during ongoing communications. The

bit synchronizer data and clock outputs are converted to optical signals

and transmitted to the indoor electronics via the A3 fiber optic cable.

Health status reporting and other test functions are eliminated from

this design. It is necessary, however, to build in certain safeguards.

Synthesizer phase-lock failure or any other condition that would cause

interference to any service must be detected and the transmitter shut

down. Since the clock signal from the bit synchronizer is normally

continuous, a simple and cost-effective clock activity monitor in

Subassembly A2 could serve as a shut-down alarm.

Page 9- 7
Subassembly A2 performs the following functions:

• All necessary automatic station control for both the indoor and

outdoor assemblies (outdoor control via fiber optic cable)

• Unique word detection, data demultiplexing (selection of

low-data-rate traffic for a particular station based on time

position with respect to the unique word)

• Decoding of received data stream

• D/A conversion for voice traffic and provision of line drivers for

data traffic.

9.2. Key Design Issues

Several aspects of the terminal design deserve special attention in

the effort to strike a proper balance between performance and cost. These

issues are discussed below.

LNA Configuration. The merits of three different receiver

configurations are compared in Table 9-I. The all-discrete approach

provides the lowest noise figure and the highest possible gain, but at the

highest cost. The completely monolithic approach provides the lowest

production cost at the expense of an unacceptably high noise figure. The

preferred approach is to realize the first stage of the LNA as a discrete

HEMT amplifier yielding 9 dB of gain and resulting in an LNA noise figure

of 2.3 riB, with the remaining gain realized by a monolithic gain block

having a noise figure of 5 dB. The g-riB gain of the first stage minimizes

the noise figure contribution of the monolithic stages.

Page 9- 8

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Table 9- I

Configuration

All discrete receiver

Discrete/MMIC receiver

All MMIC receiver

LNA Implementation Tradeoff

Development

Cost

Production Noise

Cost

Highest Lowest

Moderate Low

Lowest Highest

Lowest

High

High

LNA Filters. Losses contributed by the filter preceding the LNA

contribute significantlyto the receivernoise figure. To hold the insertion

loss to a tolerable level,a two-stage approach was adopted. A three-pole,

E-plane waveguide filterwith a 0.2 to 0.3 dB insertion loss precedes the

LNA, and a five-pole microstrip filterfollows the LNA. The latterchoice

lowers the composite filtercost and, because of its placement after the

LNA, does not significantly impact the LNA noise figure. The

LNAlbandpass filtercombination has a noise figure of 2.6 dB. However,

the low-pass filter,polarizer,and waveguide interconnects introduce 1.5

dB of loss, thereby increasing the receiver noise figure,as measured at

the antenna terminals,to 4.5 dB.

HPA Configuration. Figure g-3 shows the IMPATT amplifier design

and a recommended power profile. The amplifier requires 22 individual

parts. This number is not expected to change dramatically in the next five

years. A projected cost for these parts is $2200. Although automated

parts placement, wave soldering and automated final checkout are

employed for a major portion of the terminal, the HPA remains labor

intensive. A labor cost estimate was generated using an aggressive time

Page 9- 9
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schedule for each operation and minimum labor grades. This estimate

places the laborcost at $1800, yieldinga totalHPA cost of $4000.

9.3. Terminal Cost

Cost estimates for the units identifiedin Figure 9-I are presented in

Table 9-2. These unit costs are based on quantities of 10,000. The source

of each estimate, either TRW or a vendor, is indicated in a column at the

right of the table. The designation ASIC in the Remarks column indicates

that development of an applicationspecific integrated circuit has been

assumed for the unit inquestion.

The cost estimates are summarized in Table 9-3, in which a 10

percent profit margin has been added. The total cost per terminal is

$24,475.

While the terminal cost is distributedover a large number of units,

the HPA and LNA are major cost drivers. The cost of the HPA, in

particular,is difficultto project at this time. By contrast, at Ku-band,

the combined cost of a 2W HPA and an LNA is about $1500, compared with

$6000 as shown in Table 9-2. This disparity is largely a result of the

proliferationof small earth terminals at Ku-band, in addition to the lower

cost that normally accompanies a reduction in frequency. A similar

production history at Ka-band could drive the RF component costs below

those indicatedinTable 9-2.

Page 9- 11
OF POOR QUALITY Table 9-2. Unit Cost Estimates

_,ssy Cost Data

,Io..';omponent OescriDtlor Cost ;ource

A1 10 MHz ReI. 450 Vc, pdor

A1 A/_) Mux 200 TRW

A1 Coder& Frame Form. 300 ASIC TRW

A1 Slol _,_:o. 500 _.SIC TRW

A1 X7 150 TRW

A1 QP$K Mod 10q TRW

A1 E/O tTO MHz) 150 TRW

A1 E/O (10 MHz) 150 TRW A 1 PWB 50 TRW

A1

A1

At__ 2050

A2 CPU Control 600 ASIC TRW

A_ UW Det/DS/Decoder 250 Discrete/ASiC TRW

A2 O/E (data) 100 TRW

A2 DIE (clock) 100 TRW

A2 O/E (control) 100 TRW

A_ PWB 50 TRW

A2

A___ 1200

A3 Basic Cable (100M 100 $1.00/metar TRW

A_5 Fibers (t00M) 100 $0.'20/meter/fiber TRW

A3 Connector Assy 75 Indoor TRW

A_ Connector Assy 50 Outdoor TRW

A3

A:___ 325

A4 O/E (70 MHzt 100 TRW

A4 O/E (10 MHI) 100 TRW

A4 Amo t70 MHz'J 20 Available IC TRW

A4 Amo (10 MHz) 10 Available IC TRW

,_4 BPF SAW 15 TRW

A4 Mixer 125 TRW

A4 X200 600 Vendor

A4 BPF (_,07 GH;_) 65 Vendor

A4 Amo (2.07 GHz_ 90 MMIC TRW

A4 OEMS Band Svnth 1200 Vendor

A4 X6 650 Vendor

A4 Harmonic Mixer 350 TRW

/_4 BPF (29.5-30 GHz) 85 Vendor

A4 HPA 4000 TRW

A4 Post HPA BPF 250 Vendor

A4 Assy structure tO0 TRW

A4

A4 ,=1_ 7760

,_) Pre LNA BPF 250 Vendor

A5 LNA 2000 Discrete/ MMIC TRW

A5 Post LNA BPF 85 Vendor

_.5 Harmonic Mixer 350 TRW

A5 X4 500 Vendor

A5 BPF (_) 90 Vendor

,_5 BPF ,'B_ 90 Vendor

A5 DBPSK Demod (A_ 350 Oiscrele/ MMtC TRW

A5 DBPSK Darned (B) 350 Discrete/ MMIC TRW

A5 Baseband AGC 120 TRW _ Bit Synch 850 Available MSI TRW

A5 E/O (Clock) 150 TRW

A5 E/O tData_ 150 TRW

A_ O/E (Control I 100 TRW

A5 Control Electronics 300 TRW

A5 Baseband _;wit_:_ 30 TRW

A5 Assy structure 100 TRW

A_

A5 --• 5865

A6 LPF 250 Provides 30 GHz stop TRW

A6 Main rellector 900 TRW

A6 Subrettector 300 TRW

A6 Ant Mount 350 TRW

A6 OMT& (_p CqntroI 950 TRW

A6_• 2750

Page 9-I2

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10. SUBSCRIBER CHARGE

For purposes of computing the charge imposed on a system subscriber,

it is assumed that the user terminals are owned by the system operator.

Therefore, the subscriber charge is an all-inclusive charge that covers use

of both space and ground segments. In practice, the subscriber charge

would have two components: (1) a minimum monthly fee for access to the

system, including use of a terminal installed on the customer premises,

and (2) a usage charge based on the number of channel-minutes per month

the terminal is active. For simplicity, a common terminal activity factor

is assumed. Consequently, a uniform service charge, based on the number

of minutes of terminal activity per month, is imposed on all subscribers.

The subscriber charge is computed to provide a specified rate of

return on invested capital. A iO-year period of operation is assumed. This

is preceded by a four-year period for the development, production, and

launch of the initial satellite. The remaining useful life of the satellites

on-orbit at the conclusion of operations is accounted for by introduction

of a satellite salvage value.

By virtue of prior contractual agreements, system traffic at the start

of operations is assumed to require 5,000 64-kbps channels, or roughly

half the capacity of a single satellite. Furthermore, a 20-percent annual

traffic growth is assumed. The number of satellite channels needed to

accommodate this traffic at the start of each year of operations is shown

in Table 1O- 1.

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At the start of Year 1, a single satellite ls on-orbit. The second

satellite, provided at the start of Year 2, serves as a spare satellite until

.the channel requirement exceeds 11,250. At this polnt (Year 5), two

operational satellites plus a spare are required, so a thlrd satellite Is

launched. By the start of Year 9, three satellites are needed to handle the

offered traffic. At thls polnt a fourth satellite Is launched to serve as an

on-orblt spare.

As a minimum, the number of terminals Installed on customer

premises must equal the maximum channel demand. This minimum

terminal complement suffices only if two conditions are satisfied: (I)

each active terminal ls engaged In duplex transmission (I.e., terminals and

channels are both paired to form 64-kbps duplex circuits), and (2) at peak

satelllte loading, all terminals are actlve. With respect to the first

condition, the opposlte extreme would be for each pair of terminals to

engage In one-way or simplex transmission. Thls mode of operation would

lead to a termlnai complement whlch is twlce the maximum number of

active channels. The second condition would tend to be satisfied only if

the traffic per terminal should comprlse a substantial number of

Independent transmissions (e.g., a TI carrier). In the present instance,

with at most four Independent bit streams per termlnal (corresponding,

for example, to four 16-kbps voice channels), a significant number of

terminals wlli be Idle during the peak trafflc perlod. To allow for the

possibilities of simplex transmission and idle terminals, the number of

Installed terminals (relative to the peak channel demand) is treated

parametrically in the subscriber charge calculation.

Page 10- 3
The minimum number of terminals needed to handle the traffic that

builds up by the end of each year is shown in the last column of Table

10-I. The entries indicate the number of terminals that must be added

each year. Inthe case of the firstentry, 5,000 terminals are needed at

the start of system operations, and 1,000 more are needed to handle the

increase in peak channel activityduring the first year. Each succeeding

entry is 20 percent of the cumulative totalto that point,corresponding to

the 20 percent annual rate of trafficbuildup.

For simplicity, it is assumed that all terminals needed during the

course of a year are installed by the first of the year. This conservative

assumption is offset by the optmistic assumption that the first 6,000

terminals can be manufactured and installed in the year preceding the

start of operations.

The subscriber charge is determined from the revenue stream needed

to produce a specified internal rate of return (IRR) to the system operator.

The revenue realized during each year of system operation is proportional

to the total channel activity (i.e., number of channel-minutes of service

provided) during the year. The channel activity profile in Table 10-1, on

the other hand, represents the peak channel usage, since it has been used

to compute both the number of operational satellites and the (minimum)

number of terminals needed to support the indicated level or channel

activity. It will be assumed that the total annual channel activity is

proportional to the peak channel usage. It follows that the revenue profile

for the ten-year program is proportional to the channel activity profile

given in Table 10-1. Only the level of the revenue profile remains to be

Page I O- 4

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determined. (In computing the required annual revenue, the channel

activity in a given year is taken as the average of the start-of-year and

end-of-year values.)

The required annual revenue is divided by the peak channel demand to

obtain the required revenue on a per-channel basis. The corresponding

charge per channel-minute of actual usage is derived from the assumption

that channel usage averages 9000 minutes per month, which is equivalent

to slightly more than seven hours per business day.

A number of additional assumptions are required to perform the

service charge calculation. These are listed in Table 10-2. As indicated

in Table 10-1, a maximum of four on-orbit satellites (including a spare)

are required to handle the peak system traffic. In lieu of an expenditure

for insurance covering launch and in-orbit operations (the cost and, in

fact, the availability of which are highly uncertain at this time), the

system operator is assumed to incur the cost of manufacturing a fifth

satellite. In the most optimistic scenario, no launch or on-orbit failure

occurs and the fifth satellite is never launched. The most pessimistic

scenario considered involves a launch or on-orbit failure in Year 2,

necessitating that a replacement satellite be on-station at the start of

Year 3. An earlier failure, occurring before an on-orbit spare is available,

would result in a loss of revenue and is not considered. Neither is the

possibility of a second satellite failure, which would leave the system

without an on-orbit spare.

The schedule for satellite development and manufacture is shown in

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Figure 10-1. The development period has a duration of 2-I/2 years. The

manufacture of each satellite requires 18 months; however, successive

satellites are completed one year apart. Th minimum launch interval,

therefore, is also one year. For simplicity, the indicated launch dates are

chosen to coincide with the initial service dates for the various satellites.

All launch dates are predetermined with the exception of the one

accompanied by an asterisk. This launch takes place only if there has been

a previous failure.

Program expenditures are tabulated by year in Table 10-3. The cost

of each satellite launch is spread over three years, according to the

payment schedule required by NASA. The case shown corresponds to the

"no-failure" scenario; consequently, no launch costs are associated with

the ground spare. Satellite storage costs (two years for the third

satellite, five years for the fourth satellite, and six years for the fifth

satellite) are ignored here. The control station hardware is purchased and

integrated over a two-year period; by contrast, the control station

software is developed over a four-year period.

All expenditures except those for O&M represent depreciable capital

assets. The depreciation period for all assets is five years. In general,

the depreciation period begins with the year an asset is placed in service.

For the nonrecurring satellite costs, this is taken as the first year of

operations. For each satellite actually launched, depreciation of both the

satellite and its launch vehicle begins in the year of launch. For the

scenario in Table 10-3, the ground spare is never launched; its entire cost

is assumed to be recovered at the end of the program (i.e., after ten years

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of operations). In a similar vein, the fourth satellite launched is assumed

to have a salvage value equal to its book value after two years of

depreciation. The first three satellites are fully depreciated by program

end and therefore have zero salvage value.

Program expenditures in each category of Table 10-3 are shown as a

percentage of the total program cost in the "pie" (:hart of Figure 10-2.

Approximately 54 percent of the program cost is associated with the

space segment and 46 percent with the ground segment. (O&M costs, which

for the most part involve control station personnel, are associated with

the ground segment.) Figure 10-2 corresponds to the minimum number of

user terminals. Expenditure distributions corresponding to double or

triple the minimum number of terminals are shown in Figures 10-3 and

10-4.

The annual revenue that must be derived from each active channel is

shown in Table 10-4 for several values of IRR and for terminal profiles

equal to one, two, and three times the minimum set of values. The

equivalent per-minute charge, assuming 9000 channel-minutes of use per

month, is shown in Table 10-5.

It is instructive to compare the per-minute charge in Table 10-5 for a

64-kbps channel with the corresponding charge for switched 56-kbps

service now offered terrestrially (Reference i). The latter service is

presently offered by three vendors: AT&T, MCI/SBS, and Argo, at a

per-minute usage charge ranging from $0.30 to $0.70 for a _ circuit,

plus a monthly port charge. For monthly terminal usage ranging between

Page 1O- 10

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1000 and 4000 minutes, the port charge raises the total per-minute

charge to between $0.46 and $0.95 for the case where the usage charge is

$0,30 per minute (Argo). The corresponding range for a usage charge of

$0.70 per minute (AT&T) is $0.84 to $1.25. Although MCI/SBS has a usage

charge of only $0.45 per minute, as a result of the highest port charge its

total per-minute charge ranges from $0.72 to $1.52.

The service to which the above charges apply is limited to less than a

two-mile radius surrounding the long-distance carrier's "point of

presence" (POP) in a major city. This radius can be extended by the

purchase of Dataphone digi.tal service (DDS), which is. presently available

in about 50 percent of telephone company central offices. However, DDS is

too expensive for most customer premise equipment (CPE) customers. On

the other hand, end-to-end delivery will become available with regional

Bell operating company (RBOC) offerings of 56-kbps service. Most RBOCs

were planning to begin offering this service by the end of 1986.

For comparison with the above charges for terrestrial switched

56-kbps service, the per-minute charge in Table 10-5 must be doubled so

that it corresponds to a duplex circuit. When this is done, it is found that

the charge for satellite service is comparable to that for terrestrial

service. The former, however, would afford considerably greater

flexibility in terms of customer location.

Sensitivity of the per-minute charges in Table 10-5 to changes in the

major system cost parameters was determined. For this purpose, the

baseline case corresponds to the minimum number of user terminals and a

Page 1O- 16

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25-percent IRR. A S0-percent increase in nonrecurring satellite costs

produces a 7.2-percent increase in service charge. A 20-percent increase

in recurring satellite costs results in a 5.4-percent increase in service

charge. A virtually identical increase in service charge accompanies a

20-percent increase in launch costs. For the latter comparison, a fifth

launch (i.e., of the ground spare) is assumed to be necessary. As a result,

there is a one-to-one association of launch vehicles and satellites. Since

launch costs and recurring satellite costs are comparable, similar service

charge sensitivity to increased costs in either area can be expected.

A nondiscounted cash flow profile for the baseline case (25-percent

IRR) is shown in Figure 10-5. The solid curve shows the annual cash flow,

while the dashed curve depicts the cumulative cash flow. A similar pair

of curves, representing the discounted cash flows corresponding to a

25-percent IRR, is shown in Figure 10-6. The cumulative discounted cash

flow at program end, which by definition is the net present value for the

program, is necessarily equal to zero.

Reference:

I0-1. "The How and Where of Switched 56-kbit/s Service", Data

Communications, August 1986, pp. t 19- 133

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II. CONCLUSIONS

A satellite system architecture has been developed to pr.ovide

point-to-point voice and/or data communications among a large number of

CPS terminals. The system features narrowband FDMA uplink transmission

and wideband TDMA downlink transmission. Payload weight and power are

held to manageable levels through use of digital bulk demodulator

technology. Anticipation of advances in large-scale memory wafers has

led to the choice of a "memory-based" baseband processor to multiplex

the demodulator outputs onto the downlink carriers. Rapidly scanning

downlink beams are provided by a pair of 20-GHz array-fed dual reflector

antennas. Implementation of the feed arrays at reasonable cost depends on

MMIC technology developments.

Realization of the above system architecture at Ka-band requires the

use of coding for error correction and, in addition, provision of sizable link

margins to overcome rain attenuation. Introduction of coding limits the

capacity of a single satellite (for a 500-MHz frequency allocation) to

approximately 720 Mbps. The link margin requirements restrict the

end-to-end link availability in the worst case to 0.99. In addition, the

high cost of 30-GHz power amplifiers limits the terminal transmission

rate to about 64 kbps.

The traffic scenario adopted for computation of the user service

charge begins with a peak demand of 5,000 64-kbps channels and grows at

a 20-percent annual rate to nearly 31,000 channels after ten years of

operation. Since a single satellite has a capacity of 11,250 channels,

Page 11-I
three operational satellites (plus an on-orbit spare) are needed toward the

end of the program. In addition, production of a ground spare has been

assumed in lieu of insurance to cover launch or on-orbit failures.

The required subscriber charge depends primarily on the desired IRR

and the number of terminals in the system. (User terminals are assumed

to be owned by the system operator; consequently, terminal costs are

reflected in the subscriber charge.) For an IRR of 25 percent, for example,

the charge is in the neighborhood of $0.50 per channel-minute, the exact

amount depending on the number of terminals. This figure is comparable

to the charge for terrestrial 56-kbps switched service offered by longdistance careers.

The question of how single-hop interconnectivity is to be maintained

in a muitisatellite environment has not been addressed in this study. For a

set of terminals that need to communicate only among themselves,

single-hop interconnectivity can be established by assigning the terminals

to (i.e., pointing the terminal antennas at) a common satellite. More

generally, intersatellite links can be used to establish single-hop

interconnectivity.

Page 11-2

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i. Report No.

NASA CR 179.592

4. Title and Subtitle

2. Government Accesslon No.

Advanced Space Communications Architecture Study

Volume 2 - Technical Report

7. Author(s)

Michael Horstein, Principal Investigator

Peter 3. Hadinger, Collaborator

g. Parlormlng Organization Name and Address

TRW Inc.

One Space Park

Redondo Beach, CA 9027g

12. Sponeorlng Agency Name and Address

NASA Lewis Research Center

21000 Brookpark Road

Cleveland, OH 44135

3. Reclplent's Catalog No.

5. Report Date

March 1987

6. Psrlormlng Organization Code

8. Pertormlng Organization Report No.

10. Work Unit No.

11. Contract or Grant No.

NAS3-24743

13. Type of Report and Period Covered

Contractor Report

Jan 1986 to March 1987

14. Sponsoring Agency Code

15. Supplementary Notes

Contract Monitor - Grady H. Stevens

NASA Lewis Research Center

Cleveland, OH 44135

21 6:433-3463

16. Abstract

This study investigates the technical feasibility and economic viability of satellite

system architectures that" are suitable for customer premise service (CPS)

communications. System evaluation is performed at 30/20 GHz (Ka-band);

however, the system architectures examined are equally applicable to 14/II GHz

(Ku-band). Emphasis is placed on systems that permit low-cost user terminals.

Frequency division multiple access (FDMA) is used on the uplink, with typically

I0,000 simultaneous accesses per satellite, each of 64 kbps. Bulk demodulators

onboard the satellite, in combination with a baseband multiplexer, convert the

many narrowband uplink signals into a small number of wideband data streams for

downlink transmission. Single-hop network interconnectivity is accomplished

through use of downlink scanning beams.

Each satellite is estimated to weigh 5600 Ib and consume 6850W o_ power; the

corresponding payload totals are I000 Ib and 5000W. Nonrecurring satellite cost is

estimated at $II0 million, with the first-unit cost at $113 million. In large

quantities, the user terminal cost estimate is $25,000.

For an assumed traffic profile, the required system revenue has been computed as

a function of the internal rate of return (IRR) on invested capital. The equivalent

user charge per-minute of 64-kbps channel service has also been determined.

.

17. Key Words (Suggested by Authorls))

Satellite communications

Customer premise service

Bulk demodulators

Phased-array antennas

19. Security Classlf. (of this report)

Unclassified

20. Security Classil. (of this page)

Unclassified

18. Distribution Statement

Unclassified-Unlimited

"For sale by the National Technical Information Service, Soringfield, Virginia 22161

i21. No. of pages 22. Price"

186