# Patent

## Title
Optical waveform for use in a DWDM optical network and systems for generating and processing same

## Abstract

An optical signal occupying one or more wavelengths. An optical data signal on each wavelength is modulated with a respective overhead (dither) signal, resulting in a respective dithered optical signal. The amplitude of a particular overhead signal used to modulate the corresponding optical data signal is chosen so that the RMS value of the overhead signal in the dithered optical signal is proportional to the average intensity of the optical data signal. The instantaneous frequency of each overhead signal is time-varying and each possible frequency belongs to a distinct set of frequencies which are all harmonically related to a fundamental frequency. The distinctness of each set of frequencies allows each overhead signal to be uniquely isolated from an aggregate overhead signal. The harmonic relationship among the frequencies allows improved accuracy of RMS detection at a receiver as well as reduced computational complexity, as each possible frequency for each overhead signal can be made to fall at the center of one of the frequency bins of a single FFT of reasonable size performed at a receiver. Methods and systems for generating and detecting such signals are disclosed.

## Background
This application is a division of U.S. patent Ser. No. 09/539,706 filed Mar. 31, 2000 now U.S. Pat. No. 7,197,243.


FIELD OF THE INVENTION
The present invention relates generally to dense wavelength division multiplexed (DWDM) optical networks and, in particular, to optical waveforms used in such networks and to systems used for generating and decoding such waveforms.
BACKGROUND OF THE INVENTION
In a dense wavelength division multiplexed (DWDM) optical network, an aggregate optical signal travelling on a single optical fiber occupies multiple wavelengths, each of which carries a respective high-speed optical data signal. Although they may travel together along multiple spans throughout the network, the individual optical data signals carry information that is independent and is associated with its own source and destination network elements.
It is often desirable for a downstream network element in a DWDM network to have knowledge of the average intensity of the optical data signal occupying each wavelength. This is to allow the network element to initiate control functions such as amplifier gain adjustment and protection switching in a timely fashion. In order to keep costs to a minimum, it is desirable for the downstream network element to have access to the intensity information at multiple wavelengths without having to resort to wavelength demultiplexing and conversion of multiple (possibly hundreds of) optical signals into electrical form.
A solution to this problem for low density wavelength division multiplexed optical networks is described in U.S. Pat. No. 5,513,029 to Roberts et al., issued on Apr. 30, 1996, assigned to the assignee of the present invention and hereby incorporated by reference herein in its entirety.
In accordance with U.S. Pat. No. 5,513,029, a transmitter first modulates the optical data signal on each wavelength with a respective dither signal of a certain amplitude. The dither signals are described as being pseudo-random noise (PN) sequences which are distinct for each wavelength. The transmitter then measures the modulation depth of each individual resultant signal after modulation with the corresponding dither signal and adjusts the amplitude of the dither signal until a known “modulation depth” is achieved. (The “modulation depth” is commonly defined as the root-mean-square (RMS) value of the dither signal divided by the average intensity of the optical data signal being modulated.)
Thus, a receiver located at a downstream network element can estimate the average intensity of the optical data signal at multiple wavelengths of an incoming optically multiplexed signal by: (a) tapping a portion of the multiplexed signal using an inexpensive optical coupler; (b) converting the tapped optical signal to electrical form; (c) low-pass filtering the electrical signal to provide an aggregate dither signal; (d) separating the dither signals by matching them with known PN sequences; (e) measuring the RMS value of each dither signal; and (f) dividing the RMS value of each dither signal by the known modulation depth.
Unfortunately, the system disclosed in U.S. Pat. No. 5,513,029 has several drawbacks, which either cause the system to have poor performance or make the system prohibitively complex to implement, or both. This is especially true in the context of a dense WDM network, where there may be hundreds of optical carriers sharing a relatively narrow portion of the electromagnetic spectrum.
For example, in order to achieve good performance at the receiver, the RMS value of each dither signal present in the aggregate dither signal must be determined to a high degree of accuracy. Of course, this can only occur if the receiver accurately detects the presence of each dither signal in the aggregate dither signal. If the individual dither signals are PN sequences (as per U.S. Pat. No. 5,513,029), then in order to achieve the requisite degree of accuracy, the receiver must sample the aggregate dither signal at the center of the “chip period” of each PN sequence.
However, because the dither signals are generated on an independent basis, possibly by sources located at different parts of the network, there will generally be phase offsets among the dither signals contained in the aggregate dither signal. Therefore, in order to compensate for such differences in phase, the sampling rate used at the receiver must be much higher than the chip rate. It is therefore apparent that there exists a trade-off between sampling precision (leading to accurate RMS estimates and hence accurate estimates of average intensity) and computational simplicity.
The problem is no less severe if the dither signals used for different optical wavelengths occupy different frequencies rather than different PN sequences. In this case, the step of distinguishing the various dither signals would involve passing the aggregate dither signal through a parallel array of filters, one for each wavelength. As the number of wavelengths (and filters) increases, the bandwidth of each filter will have to be decreased in order to allow only the desired dither signal to pass through to the RMS detection stage, with the effect of deleteriously increasing the filter design complexity.
Moreover, it has been observed that framing of an optical data signal at 8 kHz causes the appearance of spurious frequency lines at multiples of 8 kHz in the frequency spectrum of the optical data signal. If such lines fall within the bandwidth of one or more of the filters, then it is apparent that the RMS value estimated as a function of each such filter's output will be biased. In a more general case, e.g., where the optical data signal is asynchronous, it may be impossible to predict the location of spurious frequency contamination, with the end result being the production of randomly biased RMS estimates.
Clearly, it would be a huge advantage to provide a system which is capable of estimating the average intensity of an optical signal at each of one or more wavelengths with reasonable accuracy and low computational complexity.
SUMMARY OF THE INVENTION
The invention is directed to providing a method and system for estimating the average intensity of an optical signal at each of one or more wavelengths with sufficiently high accuracy and sufficiently low computational complexity.
In the present invention, the optical data signal on each wavelength is modulated with a respective overhead (dither) signal, resulting in a respective dithered optical signal. The amplitude of a particular overhead signal used to modulate the corresponding optical data signal is chosen so that the RMS value of the overhead signal in the dithered optical signal is proportional to the average intensity of the optical data signal. The instantaneous frequency of each overhead signal is time-varying and each possible frequency belongs to a distinct set of frequencies which are all harmonically related to a fundamental frequency.
The distinctness of each set of frequencies allows each overhead signal to be uniquely isolated from an aggregate overhead signal. In this way, one avoids having to perform optical demultiplexing of the individual dithered optical signals at a receiver in order to extract the individual overhead signals. Furthermore, the harmonic relationship among the frequencies allows improved accuracy of RMS detection at a receiver as well as reduced computational complexity, as each possible frequency for each overhead signal can be made to fall at the center of one of the frequency bins of a single FFT of reasonable size performed at a receiver.
According to one embodiment of the invention, there is provided an optical signal with light occupying at least one wavelength, where the light on each wavelength has an intensity that is pulse modulated by a respective optical data signal and amplitude modulated by a respective overhead signal. The overhead signal has an RMS value that is a function of an average intensity of the respective optical data signal and further having an instantaneous frequency content that is time-varying. The invention further provides that the instantaneous frequency content of each overhead signal is uniquely associated with the respective wavelength and also that the instantaneous frequency content of each overhead signal is harmonically related to a common fundamental frequency.
The number of wavelengths is usually at least two and is much larger in a dense system. The RMS value of each overhead signal may be proportional to the average intensity of the respective optical data signal. Variations of the frequency content of at least one overhead signal may encode control information. The combined frequency content of each overhead signal may span a frequency range between 300 kilohertz and 1 megahertz.
According to another embodiment of the invention, there are provided a method and system for generating a dithered optical signal, including the steps of intensity modulating an optical carrier modulated data signal in accordance with an overhead signal to produce the dithered optical signal, measuring a modulation depth of the dithered optical signal and adjusting the amplitude of the overhead signal as a function of the measured modulation depth. The overhead signal has an instantaneous frequency content which is time-varying and is uniquely associated with the optical wavelength of the carrier, wherein the instantaneous frequency content is selected from a set of frequencies which are harmonically related to a common fundamental frequency.
According to still another embodiment of the invention, there is provided a method of generating first and second dithered optical signals, each of which is generated according to the above described method. In this embodiment, it is required that the frequency content of the first and second overhead signals be harmonically related to a common fundamental frequency. The first dithered optical signal may be optically coupled to the second dithered optical signal.
According to yet another embodiment of the invention, there are provided a method and system for determining the average intensity of at least one optical data signal generated in a received optical signal having been according to any method which ensures that each optical data signal occupies a respective wavelength of interest and has an intensity that is amplitude modulated by a respective overhead signal having an instantaneous frequency content that varies with time.
In this embodiment, the method includes the steps of transforming the received optical signal into an aggregate signal in electrical form, transforming the aggregate signal into a frequency domain vector and, for each wavelength of interest:

    
    
        (a) correlating the frequency-domain vector with a plurality of harmonically related frequency-domain templates to produce plural correlation results, the templates being uniquely associated with the wavelength of interest;
        (b) processing the plural correlation results to produce a candidate frequency and a candidate RMS value; and
        (c) validating the candidate frequency and the candidate RMS value.
    
    


The step of transforming the aggregate signal into a frequency-domain vector may include sampling the aggregate electrical signal and executing a fast Fourier transform (FFT) on a finite number of samples, to produce the frequency-domain vector. The method may include the intermediate steps of acquiring the samples over an integer number of blocks each having a length substantially equal to the inverse of the fundamental frequency, summing each block of windowed samples on an element-by-element basis to produce a block of summed elements and dividing each summed element by the integer number.



BRIEF DESCRIPTION OF THE DRAWINGS
These and other aspects and features of the present invention will now become apparent to those of ordinary skill in the art upon review of the accompanying description of specific embodiments of the invention in conjunction with the following drawings, in which:
FIG. 1 is a time-domain, per-wavelength depiction of a waveform that carries control information in accordance with a specific embodiment of the present invention;
FIG. 2 is a frequency plan for multiple orthogonal frequency-shift-keyed (FSK) overhead signals associated with respective optical wavelengths, according to a specific embodiment of the present invention;
FIG. 3 is a block diagram of a system used for generating a single dithered optical signal in accordance with a specific embodiment of the invention;
FIG. 4 is a block diagram of a system used for processing a multi-wavelength signal containing multiple dithered optical signals; and
FIG. 5 is a functional block diagram illustrating operation of the basic processing unit in FIG. 4.



DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
Reference is first made to FIG. 1 which shows a plurality of dithered optical signals 10, 20, 30 that are carried on respective distinct wavelengths by a common optical transmission medium (e.g., an optical fiber or silica waveguide). In an actual DWDM network, there may be up to several hundred such dithered optical signals sharing the same optical fiber. Each of the dithered optical signals 10, 20, 30 consists of light which has been twice modulated, once by a stream of data pulses and another time by an overhead signal.
Specifically, each of the dithered optical signals 10, 20, 30 is made up of a respective one of a plurality of high-speed (e.g., 10 Gbps) optical data signals 12, 22, 32, each of which is typically binary pulse modulated light that represents a logic zero when it has a low intensity and a logic one when it has a high intensity. The desired intensity of the light at the low intensity is typically the same for all the optical data signals 12, 22, 32 while the desired intensity of the light at the high intensity will be different, depending on the optical power with which the respective optical data signal is transmitted.
Each of the optical data signals 12, 22, 32 is further amplitude modulated at its source by a respective one of a plurality of low-frequency overhead signals 14, 24, 34. The amplitude of each overhead signal is chosen such that the modulation depth (i.e., the ratio of the RMS value of the overhead signal to the average intensity of the corresponding optical data signal) is equal to a known value. This allows an eventual downstream network element to obtain, inferentially, the value of the average intensity of each optical data signal simply by measuring the RMS value of the corresponding overhead signal and multiplying it by the known modulation depth.
According to the invention, each of the overhead signals 14, 24, 34 offers frequency diversity in order to provide immunity against, for instance, framing of the optical data signals at 8 kHz. To this end, the control channel employs frequency shift keying (FSK) having a relatively small number of frequency possibilities, which also keeps the receiver complexity at a relatively low level.
The frequency of each overhead signal can thus be changed on a regular basis to provide frequency diversity. Alternatively, the frequency of each overhead signal can be changed in accordance with control information, thereby causing the overhead signal to serve as a low bit rate “control channel”. The control information could include connectivity information, such as the address of the source transmitter and/or the bit rate (e.g., OC-48, OC-192) and/or format (e.g., SONET, SDH) of the corresponding optical data signal. Further examples of useful control information may be found in co-pending U.S. patent application Ser. No. 09/199,327 of Harley et al., filed Nov. 25, 1998, assigned to the assignee of the present invention and incorporated by reference herein.
Thus far, the bit rate of each control channel, the number of frequency possibilities for each overhead signal and possible values for the frequencies themselves have not been discussed. Regarding the bit rate of each control channel and the number of frequency possibilities for each overhead signal, these should be sufficiently low in order to avoid interference of a particular overhead signal with the corresponding optical data signal and also to avoid causing wild excursions of the resulting dithered optical signal.
One example of a suitable bit rate for the control channel at a given wavelength is 2 bits per second and one example of a suitable number of frequency possibilities is 8. Given these two parameters, the minimal interval between changes to the frequency content of the overhead signal will be 1.5 seconds. Of course, other bit rates and other numbers of frequency possibilities are possible.
The issue of possible frequency values is now discussed. Firstly, it should be recognized that although the various overhead signals 14, 24, 34 occupy different wavelengths, the corresponding multiple dithered optical signals 10, 20, 30 travel together on the same fiber and therefore the overhead signals will all be combined into a single aggregate overhead signal if a single optical tap coupler is used at a downstream receiver.
In order to prevent the overhead signals 14, 24, 34 from interfering with one another when they are combined into the aggregate overhead signal, it is necessary to ensure their orthogonality. This can be achieved by assigning a unique set of frequencies to each overhead signal, with each set being uniquely associated with the wavelength of the corresponding optical data signal. Thus, each overhead signal is only allowed to occupy frequencies from an assigned set.
Next, it should also be recognized that a mere random assignment of distinct frequencies for each set will result in prohibitively complex receiver equipment, insofar as sampling the overhead signals, decoding the connectivity information and estimating the average optical intensity at each wavelength are concerned. To achieve advantages related to receiver simplification, the present invention requires that all the frequencies used for all the overhead signals be harmonically related to a common fundamental frequency (denoted f0), although f0 need not itself be a member of the set of frequencies associated with any particular wavelength.
The choice of f0 is discussed in more detail at various points in the sequel and is basically influenced by the number of wavelengths sharing the same fiber, by the number of frequencies in each set of assigned frequencies, by what is considered to be a convenient FFT size (e.g., a power of 2), and by the range of possible sampling rates at the receiver.
By way of example, a suitable frequency plan for 256 wavelength channels is shown in FIG. 2, where there are taken to be 256 different frequency sets 1001, 1002, . . . , 1256, each containing 8 individual frequencies A, B, C, D, E, F, G, H. The frequencies in each set could be evenly spaced as shown, although they are only actually required to be spaced by integer multiples of f0.
As shown, the sets of frequencies 1001, 1002, . . . , 1256 can be contiguous and each adjacent pair of frequency sets is separated by a guard band 50. In order to preserve the harmonic relationship requirement of the frequencies in each set, the guard bands 50 should each have a width that is a multiple of the fundamental frequency f0. In this particular case, the width of all guard bands 50 has been chosen to be equal to twice the fundamental frequency f0. Hence, in this example, the lowest frequencies of any two adjacent channels are separated by nine times the fundamental frequency f0. It should be understood that guard bands are not required, in which case the sets of frequencies need not be contiguous (e.g., they may be interleaved).
In the example of FIG. 2, the fundamental frequency f0 has been chosen as 32.2265625 Hz, the first set of frequencies begins at 318.75 kHz, and the last set of frequencies ends at 392.94 kHz. In this particular example, therefore, the total frequency span is equal to 74.19 kHz. In general, the lowest frequency (A) in the lowest set of frequencies (1001) should be sufficiently high to avoid small signal losses due cascaded erbium-doped fiber amplifiers (EDFAs), while the highest frequency (H) in the highest set of frequencies (1256) should be sufficiently low to avoid complexity issues at the receiver side. To meet these requirements, a suitable total frequency span will desirably occupy a subset of the 300 kHz to 1 MHz range, although other ranges are possible.
Because of the required harmonic relationship of the frequencies in each set to the fundamental frequency f0, selection of the total frequency span and the number of frequency sets has a direct impact on the fundamental frequency f0 and vice versa. Ideally, for reasons of spectral efficiency, the fundamental frequency f0 should be low enough such that the corresponding total frequency span is in a desirable range, e.g., 300 kHz to 1 MHz. The actual choice of f0 within these boundaries will affect the sampling rate (denoted fR) which must be used at a receiver and will therefore depend on receiver requirements as described in further detail later on.
Generation and detection of dithered optical signals is now described. Those skilled in the art will appreciate that multiple dithered optical signals sharing the same optical fiber are usually generated at different source points in the optical network. Therefore, there exist the separate problems of (a) generating a single dithered optical signal at a source network element and (b) processing multiple dithered optical signals at a downstream (intermediate) network element. A system for generation of a single dithered optical signal will now be described with reference to FIG. 3 and a system for processing multiple dithered optical signals will be described later on with reference to FIGS. 4 and 5.
With reference to FIG. 3, therefore, there is shown a system for generating a dithered optical signal (denoted OD) on a tributary optical fiber 306 and for coupling, at a coupler 304, the dithered optical signal OD to a main optical signal arriving on an input optical fiber 302. The dithered optical signal OD occupies a wavelength of interest, while the main optical signal on the input optical fiber 302 consists of optical signal content on zero or more wavelengths with no optical signal content on the wavelength of interest.
The dithered optical signal OD on the tributary optical fiber 306 is generated by a system comprising a variable optical modulator (VOM) 310, an optical data signal generator 312 and a control module 314. Specifically, the dithered optical signal OD is generated by the variable optical modulator 310, which modulates an optical data signal (denoted OT) received from the optical data signal generator 312 as a function of an electrical overhead signal (denoted EOH) received from the control module 314.
The system of FIG. 3 could be located at a network element such as a switch or add/drop mux/demux (ADM). The optical data signal generator 312 generically represents a source of optical data, which may originate locally or which may have arrived along an arbitrary direction from another network element.
According to a specific embodiment of the invention, the electrical overhead signal EOH produced by the control module 314 causes the dithered optical signal OD produced by the VOM 310 to have the characteristics previously described with reference to FIGS. 1 and 2. That is to say, the resulting dithered optical signal OD has a modulation depth that is related to the average intensity of the optical data signal OT, while the overhead signal EOH itself has frequency variations that provide frequency diversity. One way of providing frequency diversity is to encode connectivity information into the frequency variations of the overhead signal EOH.
Also according to a specific embodiment of the invention, the possible frequencies of the overhead signal EOH are uniquely associated with the wavelength of the optical data signal OT so that knowledge of the wavelength of the dithered optical signal carrying a particular overhead signal will not be required when detecting and decoding that overhead signal at a receiver located further downstream.
In order for the dithered optical signal OD to possess these desired characteristics, the control module 314 desirably utilizes an FSK signal generator 210 or some similarly suitable generation means for generating an electrical control signal (denoted EC) which exhibits frequency variations. These frequency variations may encode connectivity information. In either case, the frequency of the control signal EC at any given instant belongs to a set of frequencies that is uniquely associated with the wavelength of the optical data signal OT.
The FSK signal generator 210 is connected to a controllable gain amplifier 220, which generates the overhead signal EOH by amplifying the control signal EC in accordance with  an electrical gain control signal (denoted EG) received from a dither processing module 230.
The value of the gain control signal EG is controlled by the dither processing module 230 so that the resulting overhead signal EOH produced by the amplifier 220 is at the appropriate amplitude, resulting in a known (or “reference”) modulation depth (denoted MD) of the dithered optical signal OD at the output of the VOM 310.
To this end, the dither processing module 230 is made up of a feedback path 240, a detection unit 250 and a comparison unit 260. The feedback path 240 is connected between the output of the VOM 310 and an input of the detection unit 250. The feedback path 240 comprises an optical tap coupler 242, a photodiode 244 and a trans-impedance amplifier 246. The optical tap coupler 242 taps a portion of the optical power of the dithered optical signal OD. The photodiode 244 converts the tapped optical signal to a photocurrent which is then converted to a voltage by the trans-impedance amplifier 246.
The trans-impedance amplifier 246 thus supplies an input of the detection unit 250 with a feedback signal (denoted EF), which is an electrical version of the dithered optical signal OD. Thus, the feedback signal EF has an overhead portion which closely matches the low rate FSK overhead signal EOH. However, the remainder of the feedback signal EF does not necessarily accurately represent the variations in the optical data signal OT because the coupler 242, photodiode 244 and/or the trans-impedance amplifier 246 may be bandwidth-limited components. Nevertheless, it is usually the case that the feedback signal EF does have a DC voltage which is representative of the average intensity of the optical data signal OT.
The detection unit 250 comprises circuitry, control logic or software for (a) determining the RMS value of the overhead portion of the feedback signal EF; (b) determining the DC voltage of the feedback signal EF; and (c) computing the quotient of the two results in order to obtain a modulation depth estimate (denoted MD′).
In order to determine the RMS value of the overhead portion of the feedback signal EF, the detection unit 250 performs a correlation between the feedback signal EF and all possible frequencies of the electrical control signal EC. To this end, the detection unit 250 may comprise an analog-to-digital converter (ADC), a plurality of time-domain correlators, a peak detector and various digital or analog low-pass filters. The complexity requirements of the detection unit 250 are relatively low because the feedback signal EF contains a single overhead signal having a limited number of frequency possibilities.
Improved correlation accuracy and further reduced complexity can be realized through synchronous correlation of the feedback signal EF with the control signal EC. This can be achieved by providing a delayed replica of the control signal EC from the output of the FSK signal generator 210 via a delay element 255. If used, the delay element 255 should introduce a delay that is approximately equal to the delay through the amplifier 220, the VOM 310, the splitter 242, the photodiode 244 and the trans-impedance amplifier 246.
In order to determine the DC voltage of the feedback signal EF, the detection unit 250 may comprise a simple capacitor or resistive-capacitive network. In order to determine the quotient of the two RMS value of the dither signal and the DC voltage of the feedback signal, thereby obtaining the modulation depth estimate MD′, the detection unit 250 may comprise a digital division module or other suitable circuitry, software and/or control logic.
The output of the detection unit 250 is thus the modulation depth estimate MD′ which is fed to the comparison unit 260. The comparison unit 260 comprises circuitry, control logic or software for comparing the received modulation depth estimate MD′ to the reference modulation depth MD. The output of the comparison unit 260 is scaled and used to modify the electrical gain control signal EG which is fed to the amplifier 220.
In operation, the control signal EC is generated by the FSK signal generator 210 and, by virtue of its frequency variations and content, the control signal EC may encode connectivity information. The control signal EC is amplified by the amplifier 220 and emerges as the overhead signal EOH. The overhead signal EOH then modulates the intensity of the optical data signal OT, resulting in the dithered optical signal OD.
The dithered optical signal OD is tapped by the coupler 242 and subsequently converted into the feedback signal EF by the photodiode 244 and the trans-impedance amplifier 246. The detector unit 250 correlates the feedback signal EF with a (delayed) version of the control signal EC in order to determine the RMS value of the overhead portion of the feedback signal EF. The DC voltage of the feedback signal EF is obtained at a capacitor output and remains unaffected by variations due to the overhead portion of the feedback signal EF.
The detector unit 250 divides the RMS value of the overhead portion of the feedback signal EF by the average DC voltage of the feedback signal EF to obtain the modulation depth estimate MD′. The comparison unit 260 then adjusts the gain control signal EG as a function of the difference between the modulation depth estimate MD′ and the reference modulation depth MD. (The modulation depth estimate MD′ may be different from the reference modulation depth MD due to temperature fluctuations, for example.)
It should be appreciated that convergence of the feedback loop in the control unit 314 is attained when the modulation depth estimate MD′ is substantially equal to the reference modulation depth MD. In this way, it is ensured that the dithered optical signal OD has been modulated at the appropriate modulation depth MD by the overhead signal EOH.
By virtue of being coupled by the coupler 304, the dithered optical signal OD occupying the wavelength of interest shares the output optical fiber 308 with other optical signals on other wavelengths. It is envisaged that these other signals may also be dithered optical signals which have been modulated by respective overhead signals that are also FSK signals. If this is indeed the case, then the FSK signals associated with different wavelengths should have the characteristics described earlier with reference to FIGS. 1 and 2.
While traveling through a portion of the network, a multi-wavelength optical signal suffers wavelength-dependent losses introduced by the optical medium itself and by various intervening cascades of erbium-doped fiber amplifiers. These losses affect the intensity of each dithered optical signal to a degree which may depend on the wavelength. However, for a given dithered optical signal, the corresponding overhead signal and the corresponding optical data signal are affected in the same way. Therefore, the ratio of the RMS value of the overhead signal to the average intensity of the optical data signal remains constant. In other words, the modulation depth of each dithered optical data signal remains unchanged, despite attenuation of the dithered optical signal.
Hence, by measuring the RMS value of each overhead signal, an intermediate network element can infer the average intensity of the corresponding optical data signal. It may also be of interest for the intermediate network element to decode connectivity information associated with one or more optical wavelengths. It would be highly desirable to perform these tasks without having to provide optical demultiplexing circuitry at the intermediate network element. Furthermore, it would be desirable to perform these tasks with reasonably low computational complexity.
To this end, FIG. 4 shows a system for estimating the average intensity of one or more optical data signals occupying respective wavelengths of an incoming multi-wavelength optical signal arriving on an optical fiber 400. The system may also be used for decoding connectivity information associated with each of these wavelengths.
It is assumed that each individual wavelength in the incoming multi-wavelength optical signal is occupied by a dithered optical signal that has been modulated with an FSK overhead signal using a system such as that shown in FIG. 3. Also, it is assumed that the modulation scheme used for multiple wavelengths is orthogonal as previously described with reference to FIGS. 1 and 2, where the instantaneous frequency of each overhead signal belongs to a distinct set of frequencies and is harmonically related to a common fundamental frequency f0.
The system in FIG. 4 comprises a data acquisition section 410 connected to a processing section 450. The data acquisition section 410 comprises an optical splitter 412 which intercepts the optical fiber 400 that carries the incoming multi-wavelength optical signal. The splitter 412 is connected in series to other components in the data acquisition section 410, including a photodiode 414, a trans-impedance amplifier 416, a low-pass filter 418, an analog-to-digital converter 420 and an averaging unit 422.
The photodiode 414 and the trans-impedance amplifier 416 can be of standard design; these components provide the low-pass filter 418 with an electrical signal (denoted EA(t)) that is representative of the aggregate light present on the optical fiber.
Because it is only necessary to detect and decode the individual overhead signals (which have a low bandwidth in comparison with the optical data signals), it is desirable to filter out high-speed variations of the electrical signal EA(t). This is achieved by the low-pass filter 418, which can be of standard design. The low-pass filter 418 ensures that the electrical signal (denoted EL(t)) at its output contains just the aggregate of the low-frequency overhead signals present on the multiple wavelengths.
Because the overhead signals are desirably limited to between 300 kHz and 1 MHz, the low-pass filter 418 can accordingly be band limited to below 1 MHz. Those skilled in the art will appreciate that the low-pass filter 418 could also be a band-pass filter with a pass band in the range of 300 kHz to 1 MHz. In any event, the filter used should have at least as wide a bandwidth as the aggregate overhead signal while filtering out variations due to high speed data.
The analog-to-digital converter (ADC) 420 can also be of standard design and comprises circuitry for sampling the signal EL(t) received from the low-pass filter 418 at a sampling rate (denoted fR) to produce a sample stream (denoted SL(n)) at its output. Because EL(t) is a band limited signal, the sampling rate fR can actually be below the so-called Nyquist rate. This technique is known as “bandpass sampling” and requires that:

    
    
        (a) fR is at least as high as, but desirably higher than, twice the total frequency span of the signal EL(t); and
        (b) fR is less than twice the minimum frequency of the signal EL(t).
    
    


In the above described example of FIG. 2, wherein the allowable frequencies of the overhead signals spanned from 318.75 kHz to 392.94 kHz and wherein f0 was set to 32.2265625 Hz, requirements (a) and (b) restrict the value for the sampling rate fR to between 148.38 kilosamples per second (ksps) and 637.5 ksps.
In addition, it has already been mentioned that the choice of the fundamental frequency f0 will dictate the choice of the sampling rate fR. In a specific sense, fR should be chosen such that:

    
    
        (c) 2N samples are produced in T0 seconds, for an integer N, where T0 is the inverse of the fundamental frequency f0; and
        (d) it is a multiple of the fundamental frequency f0.
    
    


Requirement (c) facilitates execution of a subsequent fast Fourier transform (FFT) in the processing section 450, while requirement (d) ensures that the frequency points of this ensuing FFT will fall exactly at the frequencies chosen for the overhead signals. In the above example of FIG. 2, requirements (c) and (d) further restrict the sampling rate to either 528,000 samples per second (for an FFT size of 16,384) or 264,000 samples per second (for an FFT size of 8192).
It should therefore be apparent that there is interplay between the sampling rate fR and the fundamental frequency f0. In particular, the choice of the fundamental frequency f0 directly impacts the choice of the sampling rate fR (and vice versa), just as selection of the fundamental frequency f0 had a direct impact on the choice of the total frequency span and the number of frequencies in each frequency set (and vice versa).
Considering now the averaging unit 422, this component comprises circuitry, software or control logic for buffering the sample stream SL(n) exiting the ADC 420 for the duration of an acquisition interval of Tacq seconds. The number of samples buffered during this acquisition interval of length Tacq seconds is denoted ZTOTAL and is equal to Z0×Y, where:

    
    
        Z0 is the number of samples arriving per fundamental period of T0 seconds (recalling that T0=1/f0); and
        Y is an integer greater than or equal to 1, representing the number of “blocks” per acquisition interval of Tacq seconds.
    
    


The averaging unit 422 also comprises circuitry, software and/or control logic for performing a windowing operation on the set of ZTOTAL samples buffered during Tacq seconds. In addition, the averaging unit 422 also comprises suitable circuitry, software and/or control logic for separating the windowed set of ZTOTAL samples into Y blocks of Z0 samples and for averaging the blocks on an element-by-element basis. This produces an output block of Z0 windowed and averaged samples, conveniently denoted in matrix form by [SA(1) . . . SA(Z0)].
Thus, the output of the averaging unit 422 is a block of averaged samples [SA(1) . . . SA(Z0)], one such block being produced every Tacq seconds. It should be apparent from the above that the elements of this block of averaged samples [SA(1) . . . SA(Z0)] are samples corresponding to a windowed, periodically averaged version of the aggregate overhead signal formed from the individual overhead signals associated with the various wavelengths in the incoming multi-wavelength optical signal arriving on the optical fiber 400. It should also be apparent that Z0 will be a power of two by virtue of having chosen the sampling rate fR to result in the production of 2N samples in T0 seconds for some integer N.
The averaging unit 422 feeds the block of averaged samples [SA(1) . . . SA(Z0)] to a basic processing unit 452 located at the front of the processing section 450. In addition to comprising the basic processing unit 452, the processing section 450 also comprises a plurality (M) of validation units 454A-454M connected to the basic processing unit 452. Connected to each validation unit is a respective one of a plurality of intensity estimators 456A-456M and a respective one of a plurality of decoders 458A-458M. M could equal 256 in a specific embodiment consistent with the 256 frequency sets 1001-1256 represented in FIG. 2.
The basic processing unit 452 comprises circuitry, software and/or control logic for extracting and determining the frequency content of each individual overhead signal contained in the block of averaged samples [SA(1) . . . SA(Z0)]. Of particular interest are the RMS value and the instantaneous frequency of each overhead signal as derived from the block of averaged samples [SA(1) . . . SA(Z0)]. This is because each overhead signal consists of a tone whose instantaneous frequency of the tone belongs to a restricted set of frequencies and changes with time to provide frequency diversity and, possibly, to convey control information. Also, each overhead signal (tone) has an associated RMS value that is intended to be used by the receiver in FIG. 4 for estimating the average optical intensity of the corresponding optical data signal.
Now, it should be appreciated that the occurrence of a change in the instantaneous frequency content of an overhead signal is unpredictable from the point of view of the processing section 450. In fact, it is not unlikely that during the last Tacq seconds in which an averaging operation was performed to produce the Z0 averaged samples [SA(1) . . . SA(Z0)] arriving at the basic processing unit 452, there will have been an instantaneous frequency change for one or more of the overhead signals. In such a “transitional” case, any attempt to estimate the instantaneous frequency of an affected overhead signal, as well as the RMS value of that overhead signal at that frequency, will be flawed.
To resolve this issue (and as will be described in further detail later on with reference to FIG. 5), the basic processing unit 452 produces, for each overhead signal associated with a respective optical wavelength, a respective one of a plurality of “candidate” RMS values (denoted CRMS,A, . . . ,CRMS,M) and a respective one of a plurality of “candidate” frequencies (denoted CFREQ,A, . . . ,CFREQ,M). For each overhead signal, these two respective outputs are fed along one or more control links to the respective one of the validation units 454A-454M.
Each validation units 454i, iε{A, B, . . . , M} comprises circuitry, software or control logic for processing the candidate RMS value CRMS,i and the candidate frequency CFREQ,i associated with the respective overhead signal in order to determine whether a transition has occurred in the last Tacq seconds. If so, the validation unit in question labels the corresponding candidate frequency and its associated RMS value as being invalid.
In one specific embodiment, the operation performed by each of the validation units 454A-454M may consist of determining whether the candidate RMS value associated with the respective overhead signal is above a certain pre-determined threshold. If not, it is concluded that a transition is under way and that the candidate RMS value and the corresponding candidate frequency are invalid. Of course, care should be taken to set the threshold to a value that is sufficiently low to allow for a wide dynamic range of the overhead signal. In other words, if the threshold value is set too high, a particular validation unit may consistently conclude that the candidate RMS value is too low to be valid, whereas the low value may in actuality be caused by severe, temporary, but possibly noncritical loss conditions on the fiber.
It is noted that because Tacq is much smaller than the time during which the frequency content of an overhead signal remains constant (e.g., on the order of 1.5 seconds), an invalid version of the candidate frequency and candidate RMS value will usually be followed by one or more valid versions of the candidate frequency and candidate RMS value. When a valid candidate frequency and candidate RMS value has eventually been found, the validation unit in question (say, 454p) will send a “valid RMS value” (denoted VRMS,p) to its intensity estimator 456p along one control line and a “valid frequency” (denoted VFREQ,p) to the decoder 458p along another control line.
Each of the intensity estimators 456A-456M comprises circuitry, software or control logic for dividing the valid RMS value of each overhead signal by the respective reference modulation depth MD,A-MD,M associated with that optical wavelength, thus arriving at an estimate of the average intensity of the optical data signal carried on the corresponding optical wavelength. The optical intensity determined in this manner can subsequently be processed by a controller (not shown), which may take action such as to increase the gain or initiate protection switching if the computed intensity is found to be less than a pre-determined nominal value.
Regarding the decoders 458A-458M, each such component comprises circuitry, software and/or control logic for extracting the control information, if any, encoded by the instantaneous frequency as well as the frequency variations of the respective overhead signal. Such information can include connectivity information and could be provided to a controller at the receiver, which may take switching action based upon the decoded connectivity information.
Reference is once again made to the basic processing unit 452 in FIG. 4, operation of which is now described in greater detail with additional reference to FIG. 5, wherein is shown a functional block diagram illustrating the operational steps executed by the basic processing unit 452 when processing a block of Z0 averaged samples [SA(1) . . . SA(Z0)] received from the averaging unit 422 at intervals of Tacq seconds. It should be understood that the functional elements shown in FIG. 5 could be implemented in hardware, software, firmware or any combination thereof.
At a fundamental level, the basic processing unit 452 shown in FIG. 5 implements matched filter detection in the frequency domain which gives superior performance in terms of noise removal. To this end, the first step is the provision of a fast Fourier transform (FFT) functional element 510 at the input to the basic processing unit 452. The FFT functional element 510 performs an FFT of suitable size on the block of Z0 averaged samples [SA(1) . . . SA(Z0)], received once every Tacq seconds. The result of the FFT yields a spectrum vector denoted [XA(1) . . . XA(Z0)], which shows the spectral content at each of Z0 evenly spaced frequencies.
The execution of a fast Fourier transform is simplified by virtue of having made Z0 a power of two through proper selection of the sampling rate fR and the fundamental frequency f0. In the specific examples described above, the fundamental frequency f0 was chosen to equal 32.2265625 Hz and a suitable sampling rate was found to be either 528,000 samples per second (for Z0=16,384) or 264,000 samples per second (for Z0=8192).
It is at this point that the rationale behind requiring that a harmonic relationship be maintained between the fundamental frequency f0 and each of the possible frequencies of each overhead signal as well as the sampling rate fR becomes clear. In particular, it becomes clear that by executing a single FFT (in this case a 16-K FFT or an 8-K FFT), the discrete frequency points of the resulting spectrum vector [XA(1) . . . XA(Z0)] will correspond exactly to the collection of possible frequencies for all the overhead signals.
Continuing with the functional description of the basic processing unit 452, each of a plurality of correlators 520i,j, iε{A, B, . . . , M}, jε{A, B, . . . , N}, performs a frequency domain cross-correlation between the spectrum vector [XA(1) . . . XA(Z0)] and a respective one of a plurality of spectral templates 525i,j, iε{A, B, . . . , M}, jε{A, B, . . . , N}. In this case, M is still the number of overhead signals (i.e., the number of frequency sets in the spectrum), while N is the number of frequencies per set. It should be appreciated that N may be different for each overhead signal but is in this case assumed to be constant in the interest of clarity. In the specific example described in FIG. 2, M would equal 256 and N would equal 8.
Each of the spectral templates 525i,j can be a sparse vector having a “one” at a single frequency point of interest and a “zero” elsewhere. Each correlator 520i,j may therefore perform an element-by-element multiplication of the spectrum vector [XA(1) . . . XA(Z0)] with the corresponding spectral template 525i,j, to yield a sparse vector having zeroes at all frequencies except for the frequency of the corresponding spectral template, where the value of the sparse vector at that frequency is the value of the spectrum vector [XA(1) . . . XA(Z0)] at that frequency.
Next, a plurality of inverse FFT (IFFT) functional elements 530i,j performs short IFFTs on the output of the respective correlator 520i,j. To conserve computational resources, the length of each IFFT so performed is desirably much shorter than the length of the FFT performed by the FFT functional element 510. Those skilled in the art will have little difficulty implementing a functional element which performs a short IFFT on a longer input vector.
Because the input to each IFFT functional element 530i,j is a sparse vector with at most one non-zero element, the output of each IFFT functional element 530 consists of either zeroes or samples of a time domain sinusoid (tone) at the template frequency. The result of the IFFT performed by each of the IFFT functional elements 530i,j is thus passed through a respective one of a plurality of peak detectors 540i,j in order to determine the amplitude of the resulting time-domain signal. It should be appreciated that during a change in the instantaneous frequency of an overhead signal, two or more peak detectors associated with that overhead signal may produce sinusoids of comparable amplitude.
Thus, the output of each of the peak detectors 540i,j for a common optical wavelength, i.e., for a common value of i, is fed to one of a plurality of maximum selectors 550A-550M. Each of the maximum selectors 550A-550M is operable to identify the frequency of the tone which has the highest amplitude and to compute the RMS value of that tone. Since the relationship between RMS and peak values for a sinusoid is well known, each of the maximum detectors 550A-550M may simply convert the output of the one peak detector generating the highest peak value to an RMS value.
The frequency at which the maximum selector associated with the overhead signal on a particular optical wavelength (say, maximum selector 550p) has identified the maximum amplitude is output to the corresponding validation unit 454p as the previously described candidate frequency CFREQ,p for that overhead signal while the actual peak value (converted to an RMS value) is provided to validation unit 454p as the previously described candidate RMS value CRMS,p. Each of the validation units 454A-454M then proceeds as previously described.
Other implementation options include using a different size of FFT, using a different size of IFFT, using different methods of cross-correlation, each of which may be associated with its own trade-off in terms of performance accuracy versus computational complexity. Other options include simplifying the maximum detection process by providing a simple software algorithm to run through the results of the FFT functional element 510 and isolate, for each of the M sets of frequencies, the frequency point (and the corresponding FFT value) having the largest FFT value from among the frequency points in the set.
Having considered the operation of the basic processing unit 452 with reference to FIG. 5, one of the main advantages of the invention should become clear. Specifically, because an FFT of length Z0(=T0×fR) is performed, this will produce points at frequencies that are spaced apart equally by f0 hertz. Because of the required harmonic relationship among all possible frequencies of all the overhead signals, the frequencies at which it is required to measure the RMS value fall exactly the frequency points provided by the FFT. Of course, small deviations which do not substantially affect the measurement accuracy are permissible.
Moreover, it should be appreciated that a high degree of accuracy in RMS estimation can be attained by performing a single FFT of manageable size (e.g., of length 16,384 or 8,192) every Tacq seconds. The complexity of the system is further kept in check since the IFFTs performed by the IFFT functional elements 530i,j are much shorter than the FFT performed by the FFT functional element 510i,j.
The invention also provides improved performance as a result of the specially selected frequency range of the overhead signals, which contains frequencies sufficiently high to be immune to the small-signal response of cascaded EDFA amplifiers and sufficiently low to allow low complexity at the intermediate network element where detection is required. Moreover, the fact that the overhead signals are FSK signals with frequency content that is dynamic renders the overhead signals immune to periodic interference which may arise from the transmission of frame-based optical data signals.
Of course, there are other alternative configurations which belong within the scope of the invention. For example, it should be understood that the cross-correlation and IFFT operations performed subsequent to the long 16K FFT could be executed in sequence rather than in parallel. Moreover, each overhead signal has been described as occupying only one frequency at a time, but it is possible (although less energy efficient) to attain frequency diversity and convey control information by allowing an overhead signal to occupy more than one frequency at any given instant.
Furthermore, although the above description has dealt primarily with multi-wavelength optical signals, those skilled in the art should appreciate that the invention can be applied to the generation and reception of an optical signal occupying a single carrier wavelength in the optical spectrum. In such a case, there will be only one set of frequencies associated with the overhead signal (referring to FIG. 2) and the frequencies in that set will be harmonically related to a common fundamental frequency in the above described manner.
The overhead signal will then occupy one (or possibly more) of these frequencies at a time and the actual instantaneous frequency will vary with time in order to provide frequency diversity. The time variations could be used as a control channel to transmit control information.
The fact that frequency diversity is provided allows the system to combat interference such as periodic interference due to framing of the underlying optical data signal. The fact that the frequencies in the set are harmonically related allow the receiver system to achieve good RMS measurement accuracy at a relatively low computational complexity through a suitable choice of a sampling rate and FFT size, because each possible frequency in the overhead signal can be made to fall at the center of a frequency bin of the resulting FFT.
While specific embodiments of the present invention have been described and illustrated, it will be apparent to those skilled in the art that numerous further modifications and variations can be made without departing from the scope of the invention as defined in the appended claims.

## Claims

1. A method of determining the average intensity of at least one optical data signal, each optical data signal occupying a respective wavelength of interest in a received optical signal and having an intensity that is amplitude modulated by a respective overhead signal having an instantaneous frequency content that varies with time, the method comprising:
(1) transforming the received optical signal into an aggregate signal in electrical form;
(2) transforming the aggregate signal into a frequency domain vector;
(3) for each wavelength of interest:
(a) correlating the frequency-domain vector with a plurality of harmonically related frequency-domain templates to produce plural correlation results, said templates being uniquely associated with the wavelength of interest;
(b) processing the plural correlation results to produce a candidate frequency and a candidate RMS value; and
(c) validating the candidate frequency and the candidate RMS value.

2. A method as claimed in claim 1, wherein the step of transforming the aggregate signal into a frequency-domain vector comprises:
sampling the aggregate electrical signal; and
executing a fast Fourier transform (FFT) on a finite number of samples, to produce the frequency-domain vector.

3. A method as claimed in claim 2, further comprising, between the steps of sampling and executing an FFT:
acquiring the samples over an integer number of blocks each having a length substantially equal to the inverse of the fundamental frequency;
summing each block of windowed samples on an element-by-element basis to produce a block of summed elements; and
dividing each summed element by said integer number.

4. A method as claimed in claim 3, further comprising windowing the acquired samples prior to the step of summing.

5. A method as claimed in claim 1, wherein the step of processing comprises, for each wavelength of interest, performing an inverse fast Fourier transform (IFFT) on each correlation result, performing a peak detection on the result of each IFFT and performing a maximum detection on the set of peak detection results associated with the wavelength of interest.

6. A method as claimed in claim 1, wherein the step of validating comprises verifying whether the candidate RMS value at the candidate frequency is above a pre-determined threshold.

7. A method as claimed in claim 1, further comprising estimating the average intensity of the optical data signal at each wavelength from the respective validated candidate RMS value.

8. A method as claimed in claim 7, wherein estimating comprises dividing the respective validated candidate RMS value by a respective reference modulation depth.

9. A method as claimed in claim 1, futher comprising decoding control information from the value of the validated candidate frequency.

10. A method as claimed in claim 1, further comprising decoding control information from variations in the value of the validated candidate frequency.

11. A system for determining the average intensity of at least one optical data signal, each optical data signal occupying a respective wavelength of interest in a recieved optical signal and having an intesity that is amplitude modulated by a respective overhead signal having an instantaneous frequency content that varies in time, the system comprising:
an opto-electronic conversion unit, for converting the recieved optical signal into a time-domain electrical signal;
a frequency domain transformation unit connected to the opto-electronic conversion unit, for transforming portions of the time-domain electical signal into respective frequency-domain block;
for each wavelength, a plurality of matched filters connected to the FFT module, wherein each matched filter is adapted to correlate frequency-domain blocks recieved from the frequency domain transformatin unit with a unique template representative of frequency content which is harmonically related to a common fundamental frequency;
for each wavelength , a candidate selection unit connected to the matched filters associated with that wavelength, wherein each candidate selection unit is adapted to produce one canidate frequency and one candidated RMS; and
for each wavelength, a validation unit connected to the candidate selection units associated with that wavelength, for determining whether the candidate frequency and the cadidate RMS value are valid.

12. A system for determining the average intensity of at least one optical data signal, each optical data signal occupying a respective wavelength of interest in a received optical signal and having an intensity that is amplitude modulated by a respective overhead signal having an instantaneous frequency content that varies with time, the system comprising:
means for transforming the received optical signal into an aggregate signal in electrical form;
means, connected to the transforming means, for transforming the aggregate signal into a frequency domain vector;
means, connected to the second transforming means, for correlating, for each wavelength of interest, the frequency-domain vector with a plurality of harmonically related frequency-domain templates to produce plural correlation results, said templates being uniquely associated with the wavelength of interest;
means, connected to the correlating means, for processing, for each wavelength of interest, the plural correlation results to produce a candidate frequency and a candidate RMS value; and
means, connected to the processing means, for validating, for each wavelength of interest, the candidate frequency and the candidate RMS value.