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^''^fAU O* ^ NBS TECHNICAL NOTE 1063

U.S. DEPARTMENT OF COMMERCE / National Bureau of Standards

The Characteristics of Iris-Fed

Milllmeterwave Rectangular

Microstrip Patch Antennas
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\m STANDARDS

I oo

The Characteristics of Iris-Fed ^^m\

IVIillimeterwave Rectangular Microstrip ^'^^^

Patch Antennas

D. H. Greenlee

M. Kanda

Electromagnetic Fields Division

National Engineering Laboratory

National Bureau of Standards

U.S. Department of Commerce

Boulder, Colorado 80303

D. C. Chang

Department of Electrical Engineering

University of Colorado

Boulder, Colorado 80309

\^ /

U.S. DEPARTMENT OF COMMERCE, Malcolm Baldrige, Secretary

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director

Issued October 1983
National Bureau of Standards Technical Note 1063

Natl. Bur. Stand. (U.S.), Tech. Note 1063, 50 pages (Oct. 1983)

CODEN: NBTNAE

U.S. GOVERNMENT PRINTING OFFICE

WASHINGTON: 1983

For sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, DC 20402

Price $4.00

(Add 25 percent for other than U.S. mailing)
CONTENTS

1. Introduction 1

2. A Model of the Iris-Fed Rectangular Microstrip Patch Antenna 2

3. The Experiment 7

Fabrication of the Antennas 7

Slotted Line and Far-Field Positioner 8

Experimental Setup 10

Impedance Measurement Procedure 12

4. Experimental Results 12

5. Conclusion 19

6. References 20

Appendix A 37

Appendix B 43

m

THE CHARACTERISTICS OF IRIS-FED MILLIMETERHAVE

RECTANGULAR MICROSTRIP PATCH ANTENNAS

D. H. Greenlee, M. Kanda, and D. C. Chang'*'

Electromagnetic Fields Division

National Bureau of Standards

Boulder, Colorado 80303

The fabrication of various iris-fed millimeterwave rectangular microstrip patch

antennas is described. A model is proposed to describe the iris-fed antenna. Irises

ranging in size from 15 percent of the area of the patch to the fully open waveguide are used to couple energy into the antenna. Resonance of the antenna is observed to be

insensitive to the size of the iris for irises up to 115 percent of the size of the

patch. A study is also made of the relationship of coupling to the antenna as a function

of position of the iris with respect to the transverse plane of the waveguide, the iris

always being centered with respect to the patch. In general, the antenna has a VSWR in

the waveguide feed on the order of 5:1 at resonance, except for the fully open waveguide

which gives rise to a VSWR of 2.9:1 at resonance. Far-field antenna power patterns are observed to be quite broad with H-plane beamwidths on the order of 130°. Maximum antenna

gain is seen to be 4.5 dBi with 3 dBi typical. An initial study is made of the microstrip

patch antenna fed from a longitudinal waveguide wall. Results indicate that this feed

structure is likely to prove valuable for microstrip patch antennas with coupling at least

as good as for the transverse-fed patch added to the possibility of feeding of multiple

patches from a single waveguide.

Key words: antenna gain; aperture coupling; cavity; half-power beamwidth; iris;

microstrip; millimeterwave; patch antenna; power pattern.

1. Introduction

Over the past several years the importance of the millimeterwave spectrum has grown markedly.

Systems requiring extremely broadband transmission and secure communications, and a wide variety of

radar applications are responsible for the growth in utilization of these extremely high radio

frequencies. To utilize this portion of the spectrum efficiently, appropriate antennas are a natural

requirement. The versatility of the microstrip patch antenna at microwave frequencies suggests a potential usefulness at millimeterwaves. Low cost, ease of reproduction, ruggedness, light weight,

and low profile are among the advantages this class of antennas offers at any frequency, including

millimeterwaves.

However, it has become apparent that feed structures which operate quite well at microwave

frequencies are not viable with millimeterwaves. Losses become significant in microstrip lines and

coaxial feed components are not available above about 50 GHz. For that reason, it is desirable to investigate a new feed for the microstrip patch antennas useful at millimeter wavelengths: the iris

feed. In this structure, a small hole is opened in the metallic ground plane, and energy is coupled

through that iris into the antenna.

^Department of Electrical Engineering, University of Colorado, Boulder, Colorado 80309.
The purpose of this report is to discuss briefly the model relevant to the microstrip patch

antenna and to present experimental results for iris-fed millimeterwave rectangular microstrip patch

antennas. Work was done at millimeter wavelengths rather than scaled to lower frequencies. This made

some aspects of the work much easier due to the small component sizes. Yet further reductions in

antenna size would have introduced severe tolerance problems related to fabrication and alignment of

the antennas. Also, the high frequencies made procurement of proper test equiment and sources difficult.

2. A Model of the Iris-Fed Rectangular Microstrip Patch Antenna

A microstrip device is a three-layer device consisting of a dielectric substrate sandwiched

between two layers of conducting material. Copper- teflon-copper and copper-alumina-copper sandwiches

are commonly used in practice. In figure 1 a rectangular microstrip patch antenna is illustrated.

Reference will be made throughout this report to the dimensions and coordinate system of that drawing.

Feed effects of the microstrip patch antenna used typically at microwave frequencies are based on

a traditional coaxial probe or microstrip line feed to the antenna. The present experiment, however,

has been done on microstrip patch antennas with an iris feed through the ground plane. Clearly there

are similarities in basic assumptions and final results, particularly for the far-field pattern and

dominant mode effects, if the iris is small. But certain parameters will be radically different for

the iris-fed case, particularly the input impedance of the antenna.

One simple approach to modeling the iris-fed patch antenna is illustrated in figure 2. In this

analysis the iris is treated as a window in the waveguide; the window is capacitive, inductive, or resonant depending upon the geometry. The feed susceptance, Bf, is described by a simple reactive

element. For a rectangular iris centered with respect to a transverse plane of the waveguide, as in

figure 3, the normalized susceptance is given by [1],

2 B. = -ix /a)cot2(:tD/2a) + , t^\'nV^ ^ f g 4aDcos(-n:D/2a)

J- °^,. A. In csc(i:d/2b) + (\ /aD2) x

(1 - X 2/4a2) ^ a g g

b2

^ d^

_ 8bd

_ 2b2

^ (n„d/b)K[2mt(a - d)/b]/n2 (1)

3 2 2 Tt n=l

.1/2

K(v) = r dw r K (u) du = e"^(l + 2v/ti)

'

^ V -^w

The admittance due to the unloaded patch is then described by the patch conductance and susceptance,

Gp and Bp.

A deeper investigation, however, indicates that one cannot simply describe the iris as one susceptive element and the patch as another susceptive-conductive element acting independently. The

iris will cause the excitation of higher-order modes both in the waveguide which feeds the antenna and

in the cavity-like region under the patch. These modes will not be the same as those excited by an isolated identical iris in a transverse plane of a very long section of waveguide. Furthermore, the
cavity behavior of the patch and substrate will be disrupted if the iris is very large at all, a factor not well accounted for by this model.

A more rigorous formulation of the problem is made by treating the antenna as a radiating cavity,

as in the cavity and modal expansion models [2,3], but with energy coupled into the cavity through an aperture. The antenna and feed are then modeled as in figure 4 with the sources far to the left. To represent the fields in the aperture, an equivalent magnetic current M = n x E is assumed on the

closed cavity wall and its negative on the closed waveguide aperture. Beginning with a formula based

on the conservation of reaction over the iris [4],

/ / ((eS H^) . d?) .^ =

/ / ((e! X H^) . d?) .^ , (2)

^ ^ t guide ^ ^ t cavity

A rt

where the E are tangential electric fields in the aperture and the H are the aperture magnetic

fields, it can be shown that the normalized dominant mode admittance looking into the aperture is

in -r, n r -r, .10 -I ,_. ^0 9 ^0 ' "

- "10

where

Y V mn mn

g mn.lO Y^^ V^^

is the shunt susceptance due to the waveguide side of the aperture.

b2

„ w mn ,_,

mn^tlO mn

is the shunt susceptance due to all higher order cavity modes,

b =

J / (E^ X h ) . dS , (6) mn -^ , t mn

A

where h are the magnetic mode vectors, and, for mode index, mn, Y are the characteristic wave

mn ' ' mn

1/2

admittances and V the corresponding mode voltages in the waveguide is V,q = E,Q(ab/2) where E,q

is the dominant model amplitude and ab is the waveguide cross-sectional area. By the boundary conditions at the aperture the related cavity voltage V and n = V/V,^ can be found. (In using eq (3) V need never be calculated, however, since it cancels in every term due to the normalization.)

The final term of eq (3) accounts for the dominant mode of the antenna. The antenna is said to be resonant when this term is purely real. Of course the input susceptance of the antenna will most

likely not be zero at the resonant frequency due to the susceptance of the higher order modes,

represented by B and B in figure 4. A discussion of the way in which the resonant frequency can

be experimentally determined is presented in a later chapter.
The expression for B was obtained from a variational solution for the susceptance of an aperture

in the junction of two waveguides. That susceptance can be well approximated [4] by one half of the

susceptance of a similar aperture in a long section of the first waveguide plus one half of the

susceptance due to the aperture in a long section of the second waveguide. Correspondingly, B can

be approximated by one half of the susceptance of the waveguide-antenna aperture if that aperture were located in a long section of the waveguide.

-a The b are determined by appropriate choice of E and by using the formulas,

h (x,y) = -:— [a k cos(k x)sin(k y) + mn jto|i '- X n m n

- a k sin(k x)cos(k y) 1 (7)

y m m n -^

The resonant radian frequencies w , are well -approximated by

with k = mit/h and k„ = nn/a except for the dominant mode. In that case, k.ri must be determined, m n "^ 10

For the radiating antenna, the eigenvalues k and k take on complex values of magnitude

slightly less than mn/h and nn/i but are very well approximated by these values for the non-radiating

cavity. The one exception is k,Q, which has a value of magnitude 96 percent to 98 percent of n/h. The

method by which k,Q may be found is described below [2].

^" "lO "'lO

«in = J^TY (10)

10 \ i w

Y = radiating wall admittance at x = 0, h. w '

The solution of eq (9) is found iteratively from the first two terms of a Taylor series expansion

about Tt:

k,. = (u/h) - (A /h) (11)

10 q

2(a,-h)(Tt - A ) a3

. ^ 10 P JP^ (-,2) %+l ~ (a,^h)''^ + 2a Ti - a2 - u^ "3

10 P P

A sufficiently accurate approximation of k,Q usually found by taking four iterations, that is, q

=

4. The impedance boundary condition has little effect on the higher order wave numbers which are well

approximated by k^ = (m7c/h)2 + (mi/i)'^, mn^lO.

mn Lacking a rigorous solution for the wall admittance, one approximates G to be the conductance of w

a TEM wave radiating into half-space from a narrow parallel plate waveguide of width i [4]. The wall
susceptance is approximated by the capacitance of an open microstrip circuit. The resulting equations

for the wall admittance of the rectangular microstrip patch are:

B^ = 0.01668 1^^ Eg ^^^^

Zq = 377 Q (15)

where M/t is Hammerstad's length extension formula for a microstrip transmission line [5] and e is

Schneider's effective dielectric constant for microstrip [6]

e + 0.300 j+ 0.262

^=0.412 — —

(16) ' \

-'-''' A. 0.813

Further insight into the coupling mechanism of the antenna is gained by considering both the magnetic

and electric dipole moments of the iris. Excitation of a z-directed E field in the cavity will be due

to a magnetic dipole moment co-planar with the aperture and due to an electric dipole moment normal to the aperture [7]. If the aperture is very small the guide field pattern near the aperture plane will

be very close to that for a short-circuited waveguide with field components described by

E = E.^sin sin 18)

y 10 a X

g

.^10 . ux" 2tiz" ,,q.

H = -J-—sin cos (19)

X Z,- a X 10 g

.10 ^g Ttx" . 2uz" ,_, H = J-— ^os sin . 20)

z Zin 2a a x 10 g

For a very small iris, then, neither tangential electric fields nor normal magnetic fields will exist

in the aperture plane. Instead, one would expect only an x-directed magnetic field.

As the iris becomes larger both the magnetic and electric fields will have greater and greater

fringing effects. One would expect that some of the electric field lines would bend around normal to the aperture plane and that an electric field tangent to the aperture plane would exist across the

i ri s

Coupling into the z-directed electric fields of the cavity by P , the electric dipole normal to the aperture, will be small. Rather, the stronger coupling mechanism is that due to the transverse
magnetic dipole, M , co-planar with the aperture. In the nearly short-circuited waveguide case, a magnetic field, or its equivalent magnetic current, J , exists in the aperture plane. The transverse

magnetic dipole is proportional to J and co-directional with it. It is also a function of the iris

geometry which determines the magnetic polarizabil ity of the aperture. Excitation of E under the

patch due to M will be maximized when x" = a/2, that is, when the iris is centered with respect to the X coordinate of the waveguide opening. Given the shape of the iris, the magnetic polarizabil ity

(a dyadic), a , can be determined. The magnetic dipole in the aperture is then [7]

M^ = a . H^ . (21)

t m

This will be maximized, according to eq (19), when x" = a/2.

Calculation of the Far-field Pattern

When the iris is small with respect to the size of the patch, the radiation pattern can be well

calculated. However, as the iris becomes larger, radiation from the iris itself will become

significant.

Calculation of the radiation pattern of the antenna including the iris effects can be made by

considering a radiating aperture in the plane of the patch. The fields in this aperture are the

superposition of the fringing fields at the radiating edges of the patch and the fields due to radiation from the iris. This total field in the aperture can be described by F(x', y', 0) = A(x',

y', 0)exp[-jXx', y', 0)] where the functions A and ({, describe the amplitude and phase of the aperture

field. The far- field radiation pattern due to this aperture field can be shown to be [6]

U(e,^)=2f^// F(x',y',0) (cose + -a^. i)eJ^^^'"^^^'^°^*"y'^^'"*^dx'dy' (22)

° A

where

r = distance from source point to test point

' = W ^Y h\ ^"^

s 4^ (24)

x k 6x

and

=y4f

s^= (1 - s^ - sj/'2. (26)
Given an aperture distribution F{x', y' , 0) the radiated field pattern can be calculated from eq

(21). For a large iris this function is likely to be rather complicated. For a small iris, it will

reduce to the aperture distribution of a uniform electric field oriented in the y-direction.

3. The Experiment

Fabrication of the Antennas

Fabrication of the microstrip patch antennas used in this project involved both design, including

choice of the antenna material, and actual fabrication. Because the dielectric constant of the

substrate, e , is the most critical parameter with respect to materials used, its value was selected

with care. The tolerance given for the specified value of e was of great importance since a small

tolerance-related change in e from the value used in the design would cause a significant error in

the antenna resonant frequency as compared to the predicted value. Pertinent characteristics of these

materials are presented in table 1.

Table 1. Antenna materials characteristics

Material

Copper cladding

thickness

tan 5 Thickness (t)

Substrate material

^ = 2.20 ± 0.02

157.5 g/m^

71 X 10"^ m 0.001

0.127 + 3% mm Random glass

fiber PTFE

^

= 10.5 + 0.25

39.38 g/m^

18 X 10"^ m 0.0015

0.127 ± 3% mm

Random glass

fiber PTFE

Once the desired patch size for operation at a resonant frequency in the 60 to 65 GHz range was determined, the antenna was photo! ithographically fabricated. The first step of this process was that of

forming the exposure masks. Glass plates bearing the image of the patch were then created by exposure

in a pattern generator contolled by this tape. The plates were subsequently developed in a process

similar to that used for black and white photographs.

After the copper surfaces of the antenna material was cleaned, the side to be etched was coated

with photoresist. Positive or negative process resists were used as needed, depending upon whether

the glass plates had been developed to bear a negative or a positive image of the patch. After

appropriate baking to dry the photoresist and to ensure adhesion to the copper, the material was exposed to ultraviolet light while masked by the glass plate. Those areas of negative process

photoresist exposed to the ultraviolet light underwent a chemical change so that, on application of

the developer, a hard polymer formed. Unexposed photoresist was then rinsed away by a solvent. The
opposite photochemical reaction occurred in the case of the positive photoresist with the unexposed

region being the portion polymerized on development.

The antenna material was again baked after exposure and development to ensure proper adhesion of

the protective photoresist to the copper. Etching was accomplished in a hot bath of moderately

concentrated ferric chloride, removing all but the desired copper. The patch was then cleansed of the

polymerized photoresist in a stripping solution, leaving only the clean copper patch.

The iris feed for the antenna was fabricated in a manner similar to that for the patch. The

process used was negative to that of the patch so as to etch only a small, controlled amount of copper

from the ground plane. Alignment of the iris with the patch was facilitated by special marks

generated on both the patch and iris masks. When these marks were in proper alignment, as viewed

under a microscope, the plates were taped together with the antenna sandwiched between them.

For the present purposes this process proved satisfactory in providing good iris-patch

alignment. In every case studied the iris was centered with respect to the patch. The position of

the center of the iris with respect to a target point on the patch was estimated to be within + 0.1

mm, that is, within one tenth of the smallest dimension of the smallest iris used. For studies where

even more precision is required in positioning of the iris, an apparatus enabling simultaneous

microscopic mechanical alignment of the two masks and the antenna material would be preferable.

Slotted Line and Far-Field Positioner

Beyond fabrication of the antennas, construction of two major pieces of equipment was required:

a slotted line for impedance measurements and a spherical far-field positioner for determination of

the antenna radiation patterns. Drawings of these two devices are included in appendix A.

The slotted line consisted of a section of WR 12 waveguide (3.78 x 1.89 mm, inside dimensions)

mounted in a brass baseplate and a micrometer-and-spring driven probe assembly, scaled in 0.01 mm

increments, which traveled along a dovetail track in the baseplate. The slot itself was 20 mm long

and was tapered at both ends to minimize slot radiation and generation of spurious modes. This slot

length was equivalent to 3.2Xq at 50 GHz and 4.9\q at 75 GHz, where Xg is the guide wavelength.

The 0.38 mm-wide slot was formed by first milling away the upper wall of the waveguide and

portions of the baseplate in which it was imbedded. Two very flat plates were then cut to fit into

this milled-out region with their line of contact directly above the center line of the waveguide.

These plates were then machined along that line of contact in such a way that, on affixing them again

to the baseplate, a light gap occurred between them, thus forming the slot. This procedure was not only necessitated by the small ratio of slot width to waveguide wall thickness but also added

versatility to the device by facilitating any future modification to the slot.

The electric-field probe excited a pick-up waveguide mounted on the upper traveling probe

assembly. The signal in this section of waveguide was then detected to determine the field strength

as a function of probe position.

The probe itself consisted of a 0.25-mm diameter piece of phosphor-bronze wire. One end of the

probe was forced into a short rod of plastic with the other end protruding 6.5 mm from the end of the

rod. The plastic was in turn held in a screw-like device used to move the probe in and out of the

slotted line and pick-up waveguide assembly. Vertical positioning of the probe was controlled by the

15-turns-per-cm (40-turns-per-inch) probe screw mount, thus giving better than 0.05-mm vertical

position control

8
With the best short circuit available terminating the waveguide, a copper plate clamped tightly

to the load end of the slotted line, VSVJR of the short was measured to be 20:1 or higher across the

frequency range of operation. Similarly, the measured guide wavelength was determined to be within

0.02 mm of the calculated guide wavelength, or within 0.4 percent, for all frequencies used.

Considering the difficulties involved in millimeterwave measurements and compared to the range of

values to be measured in the course of the experiment, it was decided that the slotted line had been

well constructed for the purpose at hand.

For measurements of the far-field antenna power patterns, a spherical positioner was designed and

constructed. Whereas the short wavelengths for which it was designed allowed the positioner to occupy

only a very small space, the mechanical tolerances of construction and operation of the positioner

precluded the accurate measurement of phase.

A commonly used constraint for antenna far-field measurements is that [9]

2D2/X < R (27)

D = maximum antenna aperture dimension

R = separation between transmitting and

receiving antennas.

This formula, the aperture sizes of the antennas likely to be tested and used as standard receiving

antennas, and the construction materials available were specifically considered in design of the

positioner.

The base of the positioner consisted of a 1.2-m diameter ring cut from 2.54 cm thick plywood.

This ring was cut by a band saw while the whole piece was turned on a milling machine table. Before

cutting out the center section, a 1.3-cm deep groove was scribed and milled out of the base providing

a track for azimuthal variation of the positioner.

The upright moving arm was a semicircular arc of 2.54-cm-thick plywood cut in the same way as was

the base. This arm was mounted offset from center on plastic feet designed both to travel freely in

the azimuthal positioning track and simultaneously to maintain accurate control of the arm's

position. A semicircular horizontal brace also mounted on plastic feet designed to track in the base

azimuthal positioning groove, and a vertical brace, held the upright arm perpendicular to the base.

Aluminum and plastic were used to construct the receiving antenna elevation positioning

bracket. The C-shaped bracket slid freely along the upright arm with the gap in the "C" allowing

movement past the vertical brace. A chuck holding the receiving antenna was set in a hole drilled in

an extension of the bracket. A simple brake held the entire assembly at any desired elevation. This

bracket held the pick-up antenna in a plane perpendicular to the base of the positioner and containing

the center of the base. By loosening a set screw, the chuck and receiving antenna could be oriented

for any desired polarization. The chuck clamped the receiving antenna securely so that a constant

separation was maintained between transmitting and receiving antennas for any receiving antenna

position. At the same time, the chuck could be loosened and waveguide added or removed to change the

separation between antennas.

A standard gain horn of aperture dimensions 22 x 17 mm was used throughout the experiment as the

receiving antenna. Gain of the test antenna was determined by comparison of the test antenna signal
power level with that of a standard gain horn, calibration data of that horn being supplied by the

manufacturer.

For measurements made at a separation of R = 420 nm the far-field criterion, eq (27), was met.

This was confirmed for various test antennas by moving the receiving antenna into the separation

distance R = 210 mm at which location 20 2/x = R. The normalized patterns observed at 210 mm and

420 mm were identical.

The antenna positioner was able to maintain the separation of 410 mm within 1 mm. This is far

too much variation for accurate phase measurement but the effect on power pattern measurements was

negl igible.

Elevation angle of the positioner was accurate to less than 1° when measuring at discrete

positions. However, for continuous sweeping over a given plane, this accuracy was limited by the

recording device (see below).

Azimuthal angle of the positioner was accurate to within ± 1° for the major axes. Only the major

axes (0°, 45°, 90°, etc.) were scaled since other azimuthal positions were not required. Addition of

a complete 360° azimuthal scale could be easily accomplished for any future use of the positioner.

Obviously, as with any antenna range, proper alignment of the test antenna with respect to the

range was essential. Great care was taken to center the test antenna properly and to level the

antenna range base and the test antenna ground plane in the same plane. Repeatability of patterns when

the test antenna was rotated 180° and measurement of the same peak signal level in several azimuthal

planes for the standard gain horn confirmed the accuracy of the range alignment process.

Absorbing material was appropriately placed around the range and behind the transmitting and

receiving antennas. This eliminated all known multipath and other range reflection problems.

The far-field spherical positioner was quite suitable for the measurements to be made. Errors

due to the positioner were insignificant when compared with noise and recorder errors discussed in the

next section.

Experimental Setup

After the necessary equipment, as described above, and the first antennas for testing were constructed, the setup described below and illustrated in figure 5 was assembled. This system

consisted of five sub-systems: the millimeterwave source, the test antenna, the testing devices, the

receiving system, and the output and recording devices.

The millimeter source was a klystron. Observed range of this klystron was 59.1 to 65.6 GHz with

maximum power output on the order of a few milliwatts. Amplitude and frequency drift were negligible;

even without a phase-locking system, frequency drift amounted to less than 10 MHz total frequency

deviation over a one hour period and, in most cases, less than 3 MHz over the course of any set of

measurements. Power output as monitored was also very stable, with a maximum deviation of 0.1 dB over

one hour and generally less than 0.03 dB over any set of measurements. Broadband isolators were used

to minimize loading and resonance effects on the klystron. A tuner improved the match of the system

to the klystron.

The antennas and test equipment have been previously discussed. However, the method of mounting

the antenna to the waveguide is to be noted here.

10
A short section of waveguide was filed so that its walls were very narrow at the end. With the

antenna lying on a flat surface, the waveguide was aligned with the antenna iris and soldered at the

antenna ground plane. Visible inspection of the inside of the waveguide-ground plane joint confirmed

that a good connection had been made with no holes in the joint nor solder present inside the

waveguide. The quality of this procedure was further confirmed by soldering a section of waveguide to

a piece of the antenna material from which copper had not been etched, effectively forming a short

circuit. The measured VSWR of this short circuit was 18:1, less than 1 dB below that of the best

short circuit available. The relatively high quality of the short circuit constructed in this manner

gave confidence that the solder joints of the antennas did not significantly alter the measured

antenna input parameters.

One exception to this procedure was in the case of the fully open waveguide feed. Here all the

copper was removed from the back of the antenna. The very flat dielectric-patch sheet was taped

firmly to a flat 180-mm-diameter brass flange terminating the waveguide. The dielectric material made

good contact with the flange near the waveguide opening. Some bubbles did occur between the flange

and the dielectric at distances greater than 1 cm from the waveguide opening but were observed to have

a negligible effect on the measurement. Removing the antenna from the flange and replacing it, again

smoothina out bubble near the antenna, yielded measurements repeating the original data.

The receiving system used was a phase-amplitude receiver. One channel of this receiver was connected to a 10-dB directional coupler located between the klystron and the test equipment. This

provided a reference signal to frequency-lock the receiver and to monitor the klystron power output.

The second channel of the receiver was connected to the pick-up waveguide of the slotted line during

impedance measurements and to the receiving horn mounted in the spherical positioner during far-field

power pattern measurements.

The local oscillator of the receiver generated a fundamental frequency signal in the 2 to 4 GHz range. This signal was transmitted down a coaxial cable to a mixer connected to the mill imeterwave

circuit at the slotted line or the receiving antenna on one channel and to the 10 dB directional

coupler on the other channel. The mill imeterwave signal beat against the local oscillator's

fundamental frequency, and its harmonics were generated by the mixer. The resulting signal products

were transmitted back down the coaxial cable to the receiver which locked onto the appropriate

intermediate frequency signal. This was then heterodyned to 1000 Hz and amplified, and the output of

both channels was fed to a linear-logarithmic amplifier. The power output of the klystron, sampled by

the reference channel, was monitored on the amplifier's digital display. By means of a frequency

counter the signal product frequencies just above and just below the fundamental produced by the local

oscillator were determined. From this information the klystron operating frequency was calculated

with an error of, at most, 1 kHz.

Output of the test channel of the linear-logarithmic amplifier was connected to a voltmeter for

impedance measurements and to a chart recorder for pattern measurements. Although an x-y recorder and

an elevation angle sensor on the spherical positioner would have provided more accuracy for swept

position measurements, careful sweeping of the receiving antenna assembly and repetition of the

measurements provided accuracy of better than + 5°. This accuracy was suitable for the broad

beamwidths to be measured in the experiment and was improved by comparing swept position measurement

with discrete position measurements.

Signal power level was calibrated by comparison of the test antenna signal with the peak signal

of the standard gain horn. Output power of the horn was controlled by a precision variable

11
attenuator. Settings of the attenuator were noted along with the receiving power level for both

antennas. The gain of the test antenna was then determined from this information and the

manufacturer's calibration data for the horn, having an accuracy of ± 0.25 dB. Noise on the received

signal, however, limited the accuracy of gain measurements to ± 0.5 dB.

Impedance T'leasurement Procedure

Impedance measurements using the slotted line were made by the variable attenuator method. As the probe was moved down the slot, a constant signal level was maintained at the probe output by

varying the setting of the precision attenuator. Attenuation values were noted at a sequence of peaks

and nulls. An average peak value was determined from the observed attenuator settings by first

converting the value for each peak to its linear magnitude and then averaging all such peak readings

together. The average null value was determined similarly. Finally, the square root of the ratio of

the average peak and null readings was taken to obtain the VSWR. These peak and null readings were usually within ± 0.5 dB of the average reading although in a few cases readings were observed to differ by up to ± 0.75 dB.

Due to the high sensitivity of the receiver the probe needed to be inserted only 0.05 rrni into the

slotted section and was maintained at that depth for all measurements. This amount of probe

penetration excited the pick-up waveguide sufficiently for virtually noise-free measurements yet

without adding a significant susceptive or conductive load to the slotted section.

Null positions were observed and recorded to obtain phase information. By measuring to the

accuracy of the slotted line (± 0.005 mm) the distance from a reference plane on the slotted line to the antenna ground plane, use of a reference short was unnecessary. These measurements were made with

a high-precision optical scale. Since, as previously mentioned, calculated and measured guide

wavelength were virtually identical, the phase shift of the standing-wave pattern under observation

could be determined from knowledge of these measured dimensions, along with the position of the probe

at the null and the frequency. The normalized input impedance of the antenna under test was then

calculated by the formulas [9]

R. ~ —;; o"? \

:—T7 T » l^o)

in r^cos-^ipAs) + sin^{pAs)

__ (1 - r2)cot(pAs)

in r2cot2(pAs) + 1

and r = VSWR ,

where as = null shift with respect to a short

circuit, shift toward load implies

+AS.

4. Experimental Results

Several antennas were fabricated according to the methods outlined in section 3. The iris was centered with respect to the patch in each case. Patch and iris dimensions and the size of the iris

relative to the patch and relative to the waveguide cross-sectional area are noted in table 2. The

12
resonant frequency predicted by the transmission-line model [8] and by the modal -expansion model [9]

and the percentage differences between the observed and predicted values are presented as well.

Finally, the normalized input impedance, Z^gg, of the antenna at resonance is noted.

The resonant frequency has been defined to be the frequency at which the final term of eq (3) is

purely real. This corresponds to the solution of eq (9). Experimentally, resonance was seen to occur

at a minimum in the measured VSWR, thus at a minimum in the magnitude of the reflection coefficient,

in.

Table 2. Antenna parameters.

Antenna No. 1 No. 2 No. 3 No. 4 No. 5 No. 6 No. 7 No. 8

Material

(e^) 2.20 2.20 2.20 2.20 2.20 2.20 10.5 10.5

Patch dimensions

h (mm) 1.50 1.49 1.49 1.49 1.49 1.49 1.49 1.44

H (mm) 1.25 1.23 1.23 1.23 1.23 1.23 1.23 1.12

Iris dimensions

D (mm) 0.8862 0.745 0.745 0.725 0.745 1.450 1.588 —

d (mm) 0.862 0.392 0.451 0.372 0.392 1.156 1.333 --

Observed resonant

frequency

(GHz) 63.2 63.1 62.7 62.7 63.6 63.4 62.0 60.7

Predicted resonant

frequency

(trans, line)

(GHz) 67.4 67.9 67.9 67.9 67.9 67.9 67.9 —

Error (trans.

line)

{%) -6.2 -7.1 -7.7 -7.7 -6.3 -6.6 -8.7 —

Predicted resonant

frequency

(modal expansion) 62.4 62.8 62.8 62.8 62.8 62.8 62.8

Error (modal

expansion

{%) 1.3 0.5 -0.2 -0.2 1.3 1.0 -1.3 —

Iris Area ,cy\ Patch Area

^^' 39.5 15.9 18.7 14.7 15.9 91.2 115.0 —

Iris Area ,„» Waveguide Area

10.4 4.1 4.7 3.8 4.1 23.5 29.6 100.0

z.

7— (at resonance)

^10

0.23+ 0.30+ 0.20+ 0.15+ 0.50+ 0.20+ 0.17+ 0.33+

jO.31 jO.40 jO.42 jO.32 jl.37 jO.22 jO.25 jO.OO

13
The frequency of minimum lY-jp/Y-|^Q| , and consequently that of minimum |r|, could be determined by

differentiating eq (3) with respect to frequency and setting that expression to zero. The dependence

of eq (3) on w is quite complicated and an analytical solution difficult to find. Calculation of

Y,-p/Y-]^Q for several frequencies, and from that r, provides an easier means of showing that the resonance does indeed correspond to a minimum in VSWR.

As an example, calculation is made of Y^-^/Yj^q for antenna No. 1. Dimensions of this antenna are noted in table 2. The magnetic mode vectors, h , of the antenna are defined to be those of ^ mn

eq (7). The electric field tangential aperture distribution is approximated by

with respect to the coordinate system of figure 1. Calculating the b^^ up to b33 and finding w^^ from

eq (8) for mn^lO results in the B^. noted in table 3. From eq (1) B^ is found for the given iris

dimensions and B^, as also noted in table 3, is assumed to be one-half that value. From eq (9), as described below, oj^q ^^ found. From this and bj^Q the final term of eq (3), labeled as f(bj^Q) in table

3, is calculated. The dominant mode characteristic wave admittance of the waveguide, Yj^q, is

calculated from Y,q = (1 - (f^/f) ) tZ where Zq - 377Q and f^- is the waveguide cutoff frequency.

Finally, Vin is EiA(ab/2r2 where ab is the waveguide cross-sectional area, 3.78 x 1.89 mm. From

this, Y^p/Y-j^Q is calculated as well as the corresponding reflection coefficient.

in 10

These are noted in table 3. The calculated values indicate that |r| does indeed have a minimum at 62.4 GHz, the resonant frequency predicted by the solution to eq (9).

Table 3 Calculated reflection coefficient and related parameters for antenna No. 1

Frequency Bg Bc/V^ f(bio)

1 ^10

^•n

^10

r Yin

'

^10

r'

GHz (S/m)xl0-9

-2.18

(S/m)xl0-8

2.63 220°

(Q)

60.0 -20.3 501 21.3 -86° 0.994 175° 3.37 -65° 0.793 14-9° 52.4 -20.2 -2.15 3.31 180° 487 20.5 -84° 0.989 174° 3.13 -44° 0.640 154°

63.2 -20.1 -2.15 3.12 163° 483 20.9 -85° 0.991 175° 3.43 -54° 0.721 153°

65.0 -19.8 -2.13 2.26 135° 474 21.0 -87° 0.996 175° 3.34 -71° 0.838 148°

To further illustrate the point, (Y-jp/Y^^Q)' and r' are calculated using a value B' = Bg/10.

Maintaining the same values of the other parameters used above results in a calculated input

14
admittance more in line with that actually measured. The minimum in Ir'l occurring at 62.4 GHz and

the phase variation are more pronounced in this case since B' does not have as much effect on the

calculated (Y-jp/Y-i^Q) ' as Bg does on Y^-^/Y-|^q.

Inspection of eq (3) reveals that there is more than one minimum of I Y-jp/Y-j^gl . Indeed,

differentiating eq (3) with respect to frequency, after all of the frequency dependent effects have

been included, results in an equation of fifth order in oj. Thus, there are at least two, if not

three, minima of I Y-jp/Y-j^Ql . The locations of these minima depend on the geometry of the antenna and

of the waveguide feed.

Unless the phase of Y^-^/Y-j^q has the appropriate magnitude and is rapidly varying, |r| will have

minima at the minima of lY-jp/Y^^Ql. It follows, then, that for the antennas studied it is reasonable

to assume that the resonant frequency occurs at a frequency of minimum VSWR.

This statement is not true in general for microstrip patch antennas. This is seen in the case where Y^g has approximately the same magnitude as the resonant input admittance of the antenna and the

susceptance due to the feed, Bg and B^., is non-zero. Minimum VSWR will occur at some frequency where

Bq and B^, tend to cancel the imaginary portion of the last term of eq (3), not where that term is purely real

The calculated resonant frequency was determined from the various antenna parameters, listed in table 1 and table 2, and using the transmission line model [2] and the modal expansion model [3].

This second predicted frequency is also the resonant frequency predicted by the iris feed model.

For antenna No. 1, as an example (e^ = 2.2, h = 1.50 mm, n. = 1.25 mm, t = 0.127 mm), the

calculations proceeded as follows. For the transmission line model, e^ and h were needed only to obtain a predicted resonant frequency of 67.4 GHz. For the modal expansion model, Y^ was estimated to be 0.0024+ +jO. 00456. This was obtained by inserting the appropriate antenna parameters into eqs

(13), (14), (16), and (17). A value of \q was assumed equal to 1.04 times the wavelength

corresponding to the resonant frequency. Using Aq = as a seed, eq (9) was solved by taking four

iterations of eq (12). The resulting kj^Q found from eq (11) was k-j^Q - 1939 + j85.0 m"-*- corresponding

to f^ = ck-|^Q/2p /e^ = 62.4 + j2.7 GHz. Thus the predicted resonant operating frequency of antenna No.

1 by this model is 62.4 GHz.

In figure 6 and figure 7, the VSWR and input impedance of antenna No. 1 are plotted as a function

of frequency. The input impedance, as displayed, is referenced to the ground plane of the antenna and

includes feed effects. All input impedance plots presented in this chapter are similarly referenced

to the antenna ground plane.

Inspection of figure 6 indicates that the resonant frequency of the antenna occurs at 63.2 GHz,

the frequency of minimum VSWR of the antenna. As plotted in figure 7, the normalized input impedance

of the antenna at resonance is Z^^g^ = 0.21 + j0.31. The predicted input impedance eq (3) is

(Y^-p/Yj^Q)"-'- = 0.0046 + jO.0476. It is obvious that there is a large discrepancy between the predicted

and the observed input impedances. This is due to error in the approximation of B . The calculated

value of Bq is much larger than that indicated by the measurements. If B'g and (Y-jp/Yj^Q)' are used

instead, as noted in table 3, a more reasonable predicted input impedance of 0.23 + jO.22 is

obtained. Evidently the model for the iris feed gives a good qualitative analysis of the problem but

fails to completely account for the susceptance introduced by the iris. A similar analysis holds for

the other antennas studied.

15
In figure 8 the H-plane (<t) = 0°) and E-plane ((t) = 90°) far-field power patterns of antenna No. 1 are presented for operation at 63.2 GHz. Using the standard gain horn as a reference the gain of

antenna No. 1 was measured to be 4.0 dB relative to an isotropic source (dBi), that is, 4.0 dB greater

power received by a test antenna at the peak of the power pattern than the power received as transmitted from an isotropic radiator with equal input power. Note that the peak power level

indicated by figure 8 is approximately 1.3 dBi. The figure 4.0 dBi takes into account the loss due to the 4.8:1 VSWR at this frequency. Throughout this report the power patterns are drawn and scaled as they were measured. Comments in the text correlate the measured signal level with the mismatch to give the antenna gain.

The patterns are fairly smooth and broad, as expected for the microstrip patch antenna. Small

undulations are assumed to be due to the effects of a finite ground plane (5 x 7\q at resonance for

this antenna and similar for the other antennas). The H-plane 10 dB beamwidth is observed to be 135°

while the E-plane power level is down 10 dB in the e = 90° plane. This compares to a predicted [10]

10 dB H-plane beamwidth of 166°. A virtually identical H-plane beamwidth was predicted [10] for each

of the antennas No. 2 through No. 7, yet, as will be seen, with similarly narrower observed

beamwidths.

In figure 9 the H-plane pattern for antenna No. 1 is presented for operation at 65.

4 "and 59.2

GHz. The patterns are again quite broad and smooth, as at resonance, and the gain is observed to be

4.5 dBi and 4.0 dBi respectively. One would expect a slightly higher gain for the higher frequency

since the radiating aperture is electrically larger.

For this and all antennas tested the polarization of the far-fields was linear and in the ydirection (fig. 1). The cross-polarized signal level was, in general, weak enough to be entirely

obscured by noise.

Antennas No. 2 through No. 5 were constructed with identical patch and nearly identical iris

dimensions as noted in table 2. Antennas No. 2 and No. 3 were both soldered to the waveguide section

so that the iris was centered with respect to the waveguide opening, as were all of the antennas

except No. 4 and No. 5. Antennas No. 4 and No. 5 were connected so that the iris was off-centered

along the long axis and short axis, respectively, of the waveguide but with the iris, as always,

centered with respect to the patch. This is illustrated in figure 10.

In figure 11 through figure 14 the VSWR and Smith chart input impedance loci for antennas No. 2 and No. 3 are presented. Ideally, identical results should have been observed for the two antennas.

It is seen from the VSWR curves that the resonance of antenna No. 2 occurs at 63.1 GHz and that for

antenna No. 3 at 62.7 GHz, corresponding to errors of 0.5 percent and -0.2 percent respectively in

comparison with the predicted resonant frequency. This small discrepancy is attributable to small

differences in the patch size and secondarily, to minor variations in e^. The other minima of figure

11 and figure 12--and indeed for the VSWR vs. frequency curves of antennas No. 4 through No. 7--are

due to the complicated interaction between the patch, the iris, and the waveguide, as previously

discussed. Looking ahead, one notes that there is a minimum in the VSWR vs. frequency curves for each

of the antennas No. 2 through No. 7 near the predicted resonance frequency, 62.8 GHz. Other minima occur in each of these curves but no consistency is noted between antennas. Thus it is assumed that

the resonance of the antenna occurs at that minimum near 62.8 GHz and that the other minima are due to the patch-iris-waveguide interaction.

16
If one ignores the local maxima in VSWR at 62 GHz for antenna No. 3, and instead assumes that the curve follows the dotted line drawn in that plot, similarity is readily noted between the VSWR curves

of antennas No. 2 and No. 3, with the curve of antenna No. 3 shifted downward in frequency. The input

reactance, as seen on the Smith chart plots, is also nearly identical over the range of observation.

Discrepancies can be attributed to small variations in patch size, dielectric constant, and

patch-iris and iris-waveguide alignment. This would indicate, as previously suggested, that human and

measurement errors are greater for VSWR measurements than for phase measurements.

The H- and E-plane far-field power patterns of antenna No. 3 are shown in figure 15 for operation

at 62.7 GHz. The signal level was low at this frequency of operation. Consequently, the peak-to-peak

noise level was 2 dB at maximum received signal. This was nearly triple the noise level for all other

measurements. Even with this noise level the pattern was discernible and is seen to be broad in both

planes and similar to the patterns of antenna No. 1. The small hump in the pattern in both planes is

due to the finite ground plane and, possibly, due to the antenna not being perfectly flat as mounted. Gain of antenna No. 3 at resonance is seen to be 1.5 dBi . The patterns of antenna No. 2 were similar to these, at 63.1 GHz, but even noisier due to low klystron output power.

Although not identical, the measured parameters of these two antennas are relatively close to one another. It is felt, then, that the construction of the antennas, and particularly the antennawaveguide soldering procedure, was done in a repeatable fashion. The soldering of the antenna to the

waveguide did not introduce any appreciable errors to the system as one would have expected had solder

leaked into the waveguide or had holes been left between the two pieces. Improvements would need to be made at all steps of the fabrication process to improve repeatability between antennas

significantly.

The VSWR vs. frequency curves and the Smith chart input impedance loci of antennas No. 4 and No.

5 are presented in figure 16 through figure 19. Resonance is seen to occur at 62.7 GHz and 63.6 GHz,

respectively.

The normalized resonant reactance of antenna No. 4 is jO.32 as compared to j0.40 and jO.42 for

antennas No. 2 and No. 3. This indicates that the feed of this antenna is more capacitive. Moving

the iris in the x direction away from the center of the waveguide results in less disruption of the

magnetic fields due to the iris. The result is a net reduction in stored magnetic energy causing the

antenna to be more capacitive.

Moving the iris location in the x direction also reduces the transverse magnetic dipole in the

aperture, decreasing the coupling into the antenna. This is seen in that the resonant VSWR is higher

for this antenna than for the centered iris case.

Antenna No. 5 is seen to be much more inductive than antenna No. 3 with a resonant normalized

reactance of jl.37. Moving the small iris in the y direction, but not the x direction, causes no change in the magnetic field strength over the small iris in comparison to the centered case eq

(19). Evidently there is a significant change in the total electric field near the aperture with less

stored electric energy for antenna No. 5 than for antenna No. 2 and No. 3.

The mismatch of this antenna to the waveguide was very high across the whole frequency range

observed. Coupling of energy into the antenna was so inefficient that the radiated signal was too

small for significant far-field power pattern measurement.

17
In figure 20 the far-field H- and E-plane patterns of antenna No. 4 are presented for operation

at 62.7 GHz. The patterns were similar at 63.9 GHz and at 60.5 GHz where other minima occurred in the

VSWR curve. Antenna gain at 62.7 GHz is seen to be 1.5 dBi . The H-plane 10 dB beamwidth is observed

to be 130° while the E-plane power level is down 5 dB in the e = 90° plane. Gain at 63.9 GHz was 2.0

dBi while at 60.5 GHz it was -4 dBi, significantly lower.

The VSWR vs. frequency curve of figure 21 and the related input impedance locus plotted in figure

22 indicate that the resonance of antenna No. 6 occurs at 63.4 GHz, 1.0% above the predicted resonant

frequency of 62.8 GHz. At this frequency, H- and E-plane far-field power patterns were observed as presented in figure 23. The antenna gain as measured at the peaks of the power pattern is 2.5 dBi.

The H-plane 10 dB beamwidth is 140° while the E-plane power level is down 11 dB in the e = 90°

plane. The undulations in the pattern are due not only to the finite ground plane but also indicate

some interaction between the radiation of the patch and a secondary radiation from the iris feed.

This is not unexpected since the size of the iris of this antenna is 91.2% of the size of the patch.

In figure 24 and figure 25 the VSWR and input impedance locus of antenna No. 7 are plotted.

Resonance occurs at 62.0 GHz, the center of the broad minimum in VSWR. At this frequency, 1.3X below

the predicted resonant frequency, 62.8 GHz, the gain was measured to be 3.0 dBi. The H-plane 10 dB

beamwidth is 120° while the E-plane power level is down 9 dB at e= 90°, as seen from figure 26.

The far-field power pattern for this antenna at resonance shows a dip at e = 0° as did antenna

No. 6. Again this is to be attributed to interference between radiation from the patch and radiation

from the iris. In this case the iris is even bigger, 115 percent of the size of the patch. In figure

27 the far-field power pattern of antenna No. 7 is illustrated for operation at 65.1 GHz. Off resonance there is seen to be even greater disruption of the antenna far-field pattern, particularly

in the region e = -30° to e = 0° in the H-plane. This interference effect is even more pronounced for

antenna No. 8, the antenna fed by the fully-open waveguide.

Antenna No. 8 was constructed from a material with substrate dielectric constant e^ = 10.5.

Patch size for the antenna was 1.40 x 1.12 mm. According to the transmission line model [8] the

antenna should have been resonant at 33 GHz. Instead, the antenna was observed to resonate at 60.7

GHz as indicated from the minimum in the VSWR vs. frequency curve of figure 28 and the input impedance

locus of figure 29. Investigation of smaller patches (roughly 0.7 x 0.5 mm) revealed that no resonance occurred for those patches in the 60 to 65 GHz range. Consequently, 60.7 GHz is taken to be

the dominant mode resonant frequency of this antenna. Removal of all of the copper from under the

patch radically changes the nature of the antenna. The wall admittances can no longer be accurately

predicted by eq (13) and eq (14) precluding the use of eq (9) to predict the resonant frequency.

Indeed, the antenna is no longer truly a microstrip device but only similar to microstrip.

The far-field H- and E-plane pov;er patterns of antenna No. 8 are presented in figure 30. The

smooth, single-lobed pattern again indicates that the antenna is resonant in the dominant mode rather

than a higher order mode resonance. Gain of the antenna is 0.0 dBi. Figure 31 illustrates a much

different H-plane pattern for the antenna operated off resonance. The deep dip in the center of the

pattern indicates that at this frequency, 65.3 GHz, the patch is acting more as a reflector and

scatterer of radiation from the open waveguide rather than constructively interfering as at the

resonant frequency.

Finally, in figure 32 the H-plane pattern is shown for several frequencies with the patch offcenter 0.8 mm in the +x-di recti on (figure 1) with resect to the center of the waveguide. This figure

18
clearly shows the scattering nature of the patch. The change in the patterns with frequency is due to the varying degree of excitation of the dominant mode of the patch and to the changing electrical

lengths of the antenna geometry.

5. Conclusion

A model was proposed to describe the iris-fed patch antenna based on aperture coupling to cavities. This model provided a good qualitative understanding of the iris-fed antenna although the

calculated input impedance did not correlate highly with the measured data.

Experimental analysis was made for a variety of iris-fed patch antennas. Antennas with iris

feeds of sizes ranging from 15 percent to 115 percent of the patch size typically exhibited a minmum

VSWR in the waveguide of 5:1. The fully-open waveguide feed had a significantly lower reflection

coefficient with a VSWR in the waveguide of 2.9:1 at resonance.

It was experimentally demonstrated that the resonant frequency of the antenna could be predicted

with high accuracy by the formula related to the modal expansion model even when the iris was 15 percent larger than the patch. This was not the case for the fully-open waveguide feed.

Far-field power patterns were observed for all antennas. Maximum gain for any antenna was 4.5

dBi with 2 dBi typical. Interference between the radiation from the patch and secondary radiation

from the iris feed was noted for antennas with an iris of nearly the same, or greater, size as the

patch. For the fully-open waveguide feed this interference was severe except at the resonant

frequency with the patch centered over the waveguide. For the smaller irises, and even for the iris

slightly greater in size than the patch, the far-field patterns were quite broad with H-plane 10 dB

beamwidths on the order of 130° and the power level down 10 dB in the e - 90° plane.

It has been seen, then, that the iris feed of the microstrip patch antenna is worthy of

consideration in the design of future antennas. In particular this is true for millimeterwave

microstrip patch antennas where microstrip line and coaxial feed are impractical or unavailable.

One consideration for future study is a better understanding of the 'ris. Interaction between

the iris, the patch, and the waveguide was seen to be rather complicated. With a better understanding

of the iris behavior the iris could be designed for optimal matching of the antenna to the waveguide.

Understanding of the iris behavior requires an understanding of the relationship of the position

of the iris with respect to the waveguide and with respect to the patch. The present study shows that

the best coupling occurs when the iris is centered with respect to the waveguide. The iris was always

centered under the patch for the antennas which were investigated. Future study should consider the

coupling due to the iris when it is not centered under the patch. Determination should be made of the

optimum position of the iris with respect to the patch to correlate best the fringing fields in the

iris with the dominant mode fields under the patch.

This study has also been limited to coupling from the transverse plane of the waveguide feed.

Consideration should also be made of feeding the antenna from a longitudinal waveguide wall. It is

expected that more power could be coupled into the antenna by this feed structure since, with proper

positioning, there would be both a tangential magnetic field over the aperture and a normal electric

field. The transverse plane coupling gives rise to only tangential magnetic field over the aperture

with no normal electric field to couple into the z-directed electric fields under the patch. Initial

study of this feed structure, as reported in appendix B, indicates that the longitudinal feed provides

19
at least as good coupling as that of the transverse feed. Refinements in the measurements could well

show this to be a superior feed structure with respect to the coupling of power into the antenna.

The longitudinal feed should be studied to determine the location of the patch and iris with

respect to the waveguide for optimal coupling into the antenna due to both the electric and magnetic

di poles. A sliding short circuit terminating the waveguide beyond the antenna could be used to vary

the positions of the peaks and nulls in the waveguide standing wave pattern to increase coupling into

the antenna fed in this way. Further, power could be coupled into several patches simultaneously from

one waveguide.

Finally, application of the iris feed structure to arrays of microstrip patch antennas should be

considered. Feeding of several patches should be considered. Feeding of several patches from a longitudinal wall of the waveguide is one possibility. Feeding of one patch from the waveguide and

coupling from that patch into other patches could also be considered. Incorporation of electronically

controllable phase shifters into the feed structure should also be investigated as a means of steering

the antenna radiation.

6. References

[1] Lewin, L. ; Chang, D. C. ; Kuester, E. F. Electromagnetic waves and curved structures.

Hitchin, Herts: Peter Peregrinus; 1977.

[2] Carver, K. R. ; Coffey, E. L. Theoretical investigation of the microstrip antenna. Physic,

and Sci. Lab., New Mexico State Univ., Las Cruces, Tech. Rep. PT-00929; 1979 January 23.

[3] Lo, Y. T. ; Solomon, D.; Richards, W. F. Theory and experiment on microstrip antennas. IEEE Trans. A&P, AP-27(2): 137-145; 1979 March.

[4] Harrington, R. F. Time harmonic electromagnetic fields. New York: McGraw-Hill; 1961.

[5] Hammerstad, E. 0. Equations of microstrip circuit design. Proceedings of the 5th European

Micro. Conf., 1975 September, Hamburg; pp. 268-272.

[6] Schneider, M. V. Microstrip dispersion. Proc. IEEE, 60(1): 144-146; 1972 January.

[7] Collin, R. E. Field theory of guided waves. New York: McGraw-Hill; 1960.

[8] Silver, S. (ed.) Microwave antenna theory and design. New York: Dover; 1960.

[9] Jasik, H. (ed.) Antenna engineering handbook. New York: McGraw-Hill; 1961.

[10] Derneryd, A. G. Linearly polarized microstrip antennas. IEEE Trans. A&P, AP-24(6): pp. 846-

851; 1976 November.

20
copper patch

dielectric

substrate

ground piane

ground

plane

Figure 1.

(a)

waveguide-

(b)

(a) Rectangular microstrip patch antenna.

(b) Cut-away view.

[

iB, *jB,

Figure 2. Simple input admittance model for the iris-fed patch antenna.

21
Figure 3. Centered rectangular iris terminating a rectangular waveguide.

1:n

'In ^JBnl t JB-^ Gd^ ^JB

(b)

Figure 4. (a) Aperture coupling between a rectangular waveguide and a cavity,

(b) Network model of the structure.

22
WR-12 Waveguide

Precision Variable Attenuator

\

Test Antenna Standard

Receiving

V ^•^ Antenna

Voltmeter

Recorder

Figure 5. Block diagram of the measurement system.

59 60 61 62 63 64 65 66

Frequency (GHz)

Figure 6. VSWR vs. frequency, antenna No. 1.

23
X = 1.0

X = 0.6 Xs3.0

XsO.2

X^Ohs:::

Xs-0.2

X=-3.0

x=-o.e

X = -1.0

Figure 7. Input impedance, antenna No. 1

9 = 0"

dbi '20 dbi -40 dbi

(a)

« = o'

-90"

dbi -20 dbi -40 dbi

(b)

Figure 8. Far-field power pattern, antenna No. 1, 63.2 GHz.

(a) H-plane.

(b) E-plane.

24
e = 0'

dbi -20 dbi -40 dbi

(a)

= 0-

Figure 9. H-plane far-field power pattern, antenna No. 1.

-90

(a)

(b)

65.4 GHz. 59.2 GHz.

dbi -20 dbi -40 dbi

<b)

Figure 10. Iris-waveguide orientation.

(a) Antennas Nos. 1, 2, 3, 6, 7.

(b) Antenna No. 4.

(c) Antenna No. 5.

25
VSWR ®

x:1

60 61 62 63 64 65

Frequency (GHz)

66

Figure 11. VSWR. vs. frequency, antenna No. 2.

14

12

10

VSWR ®

x:1 6

4

2

See tex

1 1

59 60 61 62 63 64 65 66

Frequency (GHz)

Figure 12. VSWR vs. frequency, antenna No. 3.

26
X = 1.0

X = 3.0

X=0.2

X =

X = -0.2

X = -0.6 X=-3.0

X = -1.0

Figure 13. Input impedance, antenna No. 2.

X = 1.0

X = 0.6 X = 3.0

X=0.2

X =

X = -0.2

X=-3.0

X = -1.0

Figure 14. Input impedance, antenna No. 3.

27
e = o'

dbi -20 dbi -40 dbi

(a)

= 0'

-90

dbi -20 dbi -40 dbi

<b)

Figure 15. Far-field power pattern, antenna No. 3, 62.7 GHz.

(a) H-plane.

(b) E-plane.

VSWR

x:1

69 60 61 62 63 64 65 66

Frequency (GHz)

Figure 16. VSWR vs. frequency, antenna No. 4.

28
VSWR

x:1

1 \ I I I L 59 60 61 62 63 64 65 66

Frequency (GHz)

Figure 17. VSWR vs. frequency, antenna No. 5.

X = 1.0

X = 0.6 Xs3.0

X=0

X =

X = -0.2

X = -0.6 X = -3.0

X=s-1.0

Figure 18. Input impedance, antenna No. 4.

29
X = 1.0

X = 0.6

60 GHz = 3.0

X=0.2

X =

X = -0.2

X = -3.0

X = -1.0

Figure 19. Input impedance, antenna No. 5.

e = o°

dbi -20 dbi -40 dbi

(a)

« = o'

dbi -20 dbi -40 dbi

(b)

Figure 20. Far-field power pattern, antenna No. 4, 62.7 GHz.

(a) H-plane.

(b) E-plane.

30
16p

14 -

12 -

10

VSWR

x:1 ^l6 -

4 -

J L J I 59 60 61 62 63 64 65 66

Frequency (GHz)

Figure 21. VSWR vs. frequency, antenna No. 6.

X = 1.0

X = 0.6 X = 3.0

X=0.2

X =

X = -0.2

X = -0.6 X=-3.0

X = -1.0

Figure 22. Input impedance, antenna No. 6.

31
= 0"

dbi -20 dbi -40 dbi

(a)

fl = 0*

-90

dbi -20 dbi -40 dbi

(b)

Figure 23. Far-field pattern, antenna No. 6, 63.4 GHz.

(a) H-plane.

(b) E-plane.

69 60 61 62 63 64 65 66

Frequency (GHz)

Figure 24. VSWR vs. frequency, antenna No. 7.

32
X = 1.0

x = o X = 3.0

X=0.2

X =

X = -0.2

X=-3.0

X = -1.0

Figure 25. Input impedance, antenna No. 7.

9 = 0"

dbi -20 dbi -40 dbi

(a)

6 = 0"

-90

dbi -20 dbi -40 dbi

(b)

Figure 26. Far-field power pattern, antenna No. 7, 62.0 GHz.

(a) H-plane.

(b) E-plane.

33
= 0"

dbi -20 dbi -40 dbl

(a)

fl = 0"

-90

dbi -20 dbj -40 dbi

(b)

Figure 27. Far-field power pattern, antenna No. 7, 65.1 GHz.

(a) H-plane.

(b) E-plane.

lOp

8

VSWR 6

x: 1 .

± 1

59 60 61 62 63 64

Frequency (GHz)

65 66

Figure 28. VSWR vs. frequency, antenna No. 8.

34
X = 1.0

X = 0.6 X = 3.0

X=0.2

X =

X = -0.2

X=-3.0

X = -1.0

Figure 29. Input impedance, antenna No. 8.

e=:0'

dbi -20 dbi -40 dbi

(a)

6 = 0"

-90

Odbl -20 dbi -40 dbi

(b)

Figure 30. Far-field power pattern, antenna No. 8, 60.7 GHz,

(a) H-plane.

(b) E-plane.

35
« = 0"

Figure 31. H-plane far- field power pattern,

antenna No. 8, 65.3 GHz.

dbi -20 dbi -40 dbi

(a)

« = o'

-90*

dbi -20 dbi -40 dbi

(a)

e = o'

Odbi

Figure 32. H-plane far-field power pattern,

antenna No. 8, off-centered 0.8 mm in + x-di recti on.

(a)

(b)

(c)

60.7 GHz. 63.4 GHz.

65.3 GHz.

e = o"

dbi -20 dbi -40 dbi

(c)

36
Appendix A

Far-Field Positioner and Slotted Line Drawings

On the following pages drawings of the far-field spherical positioner (fig. Al through fig. A3)

and of the slotted line (fig. A4 through fig. A9) are presented. Certain non-critical dimensions have

been omitted such as dimensions for set screw holes.

I

37
Figure Al. Baseplate of spherical positioner.

9.53 mm 25.4 mm

3/8" r

i ]_

Section AA

Delrin Feet25.4 mm

Figure A2. Elevation position arm of

spherical positioner.

25.4 mm

1"

'Deh-in Feet. Center offset 1/2'

from center of upright arm. 25.4 mm

38
3* 9^3 mm, q- Bore Set screw for brake

9.53 mm

11.11mm

Figure A3. Receiving antenna mounting bracket for spherical positioner.

!

12.70 mm

.5-

41.28 mm

1.625'

101.60 mm

4.00'

107.95 mm

4.25'

3

31.75mm

1.25*

T^

A.S2

3.94 rrmi .155'

mm

" .187' 3 9.53 mm

.375" 18.75 rnm

.738' h

37.47 mm

1.475'

Cut recess after soldering in WR-12 waveguide

25.4 mm

1.00'

Figure A4. Slotted line baseplate,

39
T

6.66

8.66 mm

.341" n—rm TT .4-'U I JL

T— I—I—irr Jlh\

76.2 mm

3.0' 25.4 mm ^

1.0" 19.86 mm

.782" 5.82 mm/.229" -—^ -^ H

&-

8.66 mm

4.75 mm

.187' -*

7.1 1 mm

.280" 18.05 mm ^ -75"

5.03 mm

.198"

4.75 mm

.187"

^ 31.75mm ^ i^^_ ^ _ _ -^ w

44/45 mm

1.75"

12.7 mm

-^-60" Bore

.312"

Figure A5. Slotted line sliding section.

:zj

steel Ball

Spring Mount

Mounting Screws <|>i I iq)

I I I

.375/^9.63 mm

.186/^4.70 mm

Figure A6. Slider pusher assembly,

40
Set screws for

micrometer head

~"—'

' """

*"0

r

L n

1

< -'

14.99 mm

.59'

Dimensions as for Fig. A6 except as noted.

oil no

Figure A7. Micrometer mount for slotted line.

i

f

20.62 mm

812' _^ 7.92 mm " . ^

.312' I

9 1 mm I

154' ' -^ (-<- 6.82mm/<229'

3.18 mm 6.35 mm

.067'

center to termination

Figure A8. Pick-up waveguide termination and probe mount sleeve.

41
Set screw for probe holder

Tapped — -40

3J8^ Drill

12.7 mm

.5"

(a)

T

25.4 mm

1.00"

4.78 mm

.188"

(b)

Figure A9. (a) Probe adjustment screw,

(b) Kel-f probe holder.

42
Appendix B

Initial Investigation of the Longitudinal Feed

An initial investigation was made into the iris-fed rectangular microstrip patch antenna fed from

the longitudinal wall of the source waveguide. The patch, constructed on the material (e = 2.20), was

identical in size to the patches of antennas No. 2 through No. 7. Thus, the predicted resonant frequency was 62.8 GHz eq (21). The iris, centered with respect to the patch, was of dimensions D

=

0.372 mm and d ^ 0.725 mm.

One of the two wider walls of the WR-12 waveguide was milled away. In its place the antenna was soldered with the iris off-center 0.3 mm in the x"-direction (figure 3) from the center of the waveguide wall. The patch was oriented so that the y-direction of the patch coincided with the z"-direction of the waveguide.

With a reflectionless termination placed after the antenna section, the input impedance of the

waveguide-antenna section was measured, as plotted in figure Bl. At the predicted resonant frequency

of the antenna, 62.8 GHz, the normalized input impedance of the section was 0.87-J0.16.

The far-field power patterns were also observed at 62.8 GHz, as illustrated in figure B2. Gain

of the antenna, as compared to the FXR standard gain horn, was 4.5 dBi . Undulations in the patterns

are assumed to be due to reflectins from the test equipment. The sharp dip in the E-plane pattern at

= -90° is due to blockage from the waveguide circuit. Whereas for the transverse case the test equipment could be effectively screened by absorbing material below the antenna ground plane, for the

longitudinal feed such screening is not so readily accomplished. Future work should be done using a waveguide feed circuit so designed as to eliminate such reflection problems.

The information presented here represents only preliminary results for the longitudinally fed antenna. These results do, however, indicate promise for this feed structure. Further comments

concerning the direction of future study are made in the conclusion.

43
X = 1.0

x=o X=:3.0

X = 0.2

X =

X = -0.2

X = -0.

X = -1.0

Figure Bl. Input impedance of the longitudinally fed antenna,

dbi -20 dbi -40 dbi

(a)

= 0"

dbi -20 dbi -40 dbi

(b)

Figure B2. Far-field power pattern, longitudinally fed antenna, 62.8 GHz.

(a) H-plane.

(b) E-plane.

44
NBS-n4A IREV. 2-8C) U.S. DEPT. OF COMM. BIBLIOGRAPHIC DATA

SHEET (See instructions)

1. PUBLICATION OR

REPORT NO. NBS-TN-1063

2. Performing Organ. Report No. 3. Publication Date

October 1983

4. TITLE AND SUBTITLE

The Characteristics of Iris-Fed Millimeterwave Rectangular MicrOstrip

Patch Antennas

5. AUTHOR(S)

D. H. Greenlee, M. Kanda, D. C. Chang

6. PERFORMING ORGANIZATION ('/f joint or other thon NBS. see instructions)

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

WASHINGTON, D.C. 20234

7. Contract/Grant No.

8. Type of Report & Period Covered

9. SPONSORING ORGANIZATION NAME AND COMPLETE ADDRESS (Street. City. State. ZIP)

Same (#6)

10. SUPPLEMENTARY NOTES

I I Document describes a computer program; SF-185, FlPS Software Summary, is attached.

11. ABSTRACT (A 200-word or less factual sumnnary of most si gnificant information. If document includes a si gnificant

bibliography or literature survey, mention it here)

The fabrication of various iris- fed millimeterwave rectangular microstrip pat

antennas is described. A model is proposed to describe the iris-fed antenna,

ranging in size from 15 percent of the area of the patch to the fully open Wa

are used to couple energy into the antenna. Resonance of the antenna is obso

be insensitive to the size of the iris for irises up to 115 percent of the si

the patch. A study is also made of the relationship of coupling to the anten

function of the position of the iris with respect to the transverse plane of

waveguide, the iris always being centered with respect to the patch. In gene

the antenna has a VSWR in the waveguide feed on the order of 5:1 at resonanc

for the fully open waveguide which gives rise to a VSWR of 2.9:1 at resonance

field antenna power patterns are observed to be quite broad with H-plane beam

on the order of 130°. Maximum antenna gain is seen to be 4.5 dBi with 3 dBi

An initial study is made of the microstrip patch antenna fed from a longitudi

waveguide wall. Results indicate that this feed structure is likely to prove

for microstrip patch antennas with coupling at least as good as for the trans

patch added to the possibility of feeding of multiple patches from a single w

ch

Irises

vequide

rved to ze of

na as a the

ral,, c, except

Fa r- ividths

typical

nal

valuable

verse-fed

avoouide

12. KEY WORDS (Six to twelve entries; alphabetical order; capitalize only proper names; and separate key words by semicolons)

antenna gain; aperture coupling; cavity; half-power beamwidth; iris; microstrip;

millimeterwave; patch antenna; power pattern.

13. AVAILABILITY

[^ Unlimited

I I For Official Distribution, Do Not Release to NTIS

^3! Order From Superintendent of Documents, U.S. Government Printing Office, Washington, D.C

20402.

\2Z\ Order From National Technical Information Service (NTIS), Springfield, VA. 22161

14. NO. OF

PRINTED PAGES

50

15. Price

$4.00

•a U.S. Goverrunent Printing Office 1984 - 776-660/38 Reg. 8 USCOMM-DC 6043-PeO



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