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/ mifs'NBS TECHNICAL NOTE 1008

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U.S. DEPARTMENT OF COMMERCE/ National Bureau of Standards
NATIONAL BUREAU OF STANDARDS

The National Bureau of Standards' was established by an act of Congress March 3, 1901. The

Bureau's overall goal is to strengthen and advance the Nation's science and technology and

facilitate their effective appHcation for public benefit. To this end, the Bureau conducts

research and provides: (1) a basis for the Nation's physical measurement system, (2) scientific and technological services for industry and government, (3) a technical basis for equity in trade, and (4) technical services to promote public safety. The Bureau's technical work is performed by the National Measurement Laboratory, the National Engineering Laboratory,

and the Institute for Computer Sciences and Technology.

THE NATIONAL MEASUREMENT LABORATORY provides the national system of

physical and chemical and materials measurement; coordinates the system with measurement

systems of other nations and furnishes essential services leading to accurate and uniform

physical and chemical measurement throughout the Nation's scientific community, industry,

and commerce; conducts materials research leading to improved methods of measurement,

standards, and data on the properties of materials needed by industry, commerce, educational

institutions, and Government; provides advisory and research services to other Government

Agencies; develops, produces, and distributes Standard Reference Materials; and provides

calibration services. The Laboratory consists of the following centers:

Absolute Physical Quantities^ — Radiation Research — Thermodynamics and

Molecular Science — Analytical Chemistry — Materials Science. THE NATIONAL ENGINEERING LABORATORY provides technology and technical

services to users in the public and private sectors to address national needs and to solve national problems in the public interest; conducts research in engineering and applied science

in support of objectives in these efforts; builds and maintains competence in the necessary

disciplines required to carry out this research and technical service; develops engineering data and measurement capabilities; provides engineering measurement traceability services; develops test methods and proposes engineering standards and code changes; develops and

proposes new engineering practices; and develops and improves mechanisms to transfer

results of its research to the utlimate user. The Laboratory consists of the following centers:

Applied Mathematics — Electronics and Electrical Engineering^ — Mechanical

Engineering and Process Technology^ — Building Technology — Fire Research —

Consumer Product Technology — Field Methods.

THE INSTITUTE FOR COMPUTER SCIENCES AND TECHNOLOGY conducts

research and provides scientific and technical services to aid Federal Agencies in the selection, acquisition, application, and use of computer technology to improve effectiveness and

economy in Government operations in accordance with Public Law 89-306 (40 U.S.C. 759), relevant Executive Orders, and other directives; carries out this mission by managing the Federal Information Processing Standards Program, developing Federal ADP standards

guidelines, and managing Federal participation in ADP voluntary standardization activities; provides scientific and technological advisory services and assistance to Federal Agencies; and

provides the technical foundation for computer-related policies of the Federal Government.

The Institute consists of the following divisions:

Systems and Software — Computer Systems Engineering — Information Technology.

'Headquarters and Laboratories at Gaithersburg, Maryland, unless otherwise noted; mailing address Washington,D.C. 20234. 'Some divisions within the center are located at Boulder, Colorado, 80303.

The National Bureau of Standards was reorganized, effective April 9, 1978.
p^.ANTENNAS AND THE ASSOCIATED

TIME DOMAIN RANGE FOR THE

National BureaM ?* •''.anslarcts JAN 2 5 ITO

MEASUREMENT OF IMPULSIVE FIELDS

'

Robert A. Lawton

Arthur R. Ondrejka

Electromagnetic Technology Division

National Engineering Laboratory

National Bureau of Standards

Boulder, Colorado 80303

\

•"•CAU 0» *

U.S. DEPARTMENT OF COMMERCE, Juanita M. Kreps, Secretary

Sidney Harman, Under Secretary

Jordan J. Baruch, Assistant Secretary for Science and Technology

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director

Issued November 1978
NATIONAL BUREAU OF STANDARDS TECHNICAL NOTE 1008

Nat. Bur. Stand. (U.S.), Tech. Note 1008, 68 pages (Nov. 1978)

CODEN: NBTNAE

U.S. GOVERNMENT PRINTING OFFICE WASHINGTON: 1978

For sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, D.C. 20402

Stock No. 003-003-01998-4 Price $2.40 (Add 25 percent additional for other than U.S. mailing)
CONTENTS

Page

INTRODUCTION 1 FIELD MEASUREMENT EQUATIONS 3 MEASUREMENT SYSTEM 9 3.1 Sampling Oscilloscope 11 3.2 Computer, Peripherals, and Analog Converters 13 ANTENNA SYSTEMS 15 4.1 Standard Cone Antenna 15 4.2 TEM Horn Antenna 19 4.3 Resistively Loaded Monopole 31 GENERATORS 34 MEASUREMENT RESULTS 37

6.1 Description of Measurements 37 6.2 Evaluation of Errors 43

RECOMMENDED USE OF ANTENNAS FOR FIELD GENERATION 46

CONCLUSIONS 48

APPENDIX A 5

1

APPENDIX B 60

REFERENCES 62

iii

ANTENNAS AND THE ASSOCIATED TIME DOMAIN RANGE FOR THE MEASUREMENT OF IMPULSIVE FIELDS

Robert A. Lawton and Arthur R. Ondrejka

Electromagnetic Technology Division

Center for Electronics and Electrical Engineering

National Bureau of Standards

Boulder, Colorado, 80303 U.S.A.

This report describes the construction and evaluation of a TEM horn antenna

designed at NBS to be used as a transfer standard to generate and measure impulsive

electromagnetic fields. Our purpose in the evaluation was to analyze the different

electrical field generation and measurement techniques thoroughly enough to determine

the major sources of error and establish a standard of impulsive field strength

having a well established statement of accuracy.

The evaluation of this horn was done in two independent ways; by placing the

horn in the field of a conical transmission line and by a three antenna intercomparison. The two methods were found to agree within ± 3 dB over the range of

0.6 to 5 GHz. Part of this disagreement is due to the assumption of far field

conditions, and an experimental technique is described which determines the

frequency range over which this assumption is valid.

Key words: Antenna; conical transmission line; impulsive fields; impulse

response; standards; TEM horn antenna.

1. INTRODUCTION

Radar and its many applications for the military have been with us for some time now

while civilian applications (other than speed traps), such as automobile collision avoidance

and air bag inflation, have awaited the higher resolution that present state-of-the-art

radars are just now making available. This development, together with digital communication

systems which are now approaching the gigabit data rate, requires wideband antennas in order

to achieve the required transient response.

Present techniques for evaluating the pertinent antenna parameters include:

1. Stepping or sweeping a cw generator and observing the response through identical

transmitting and receiving antennas with a swept (or stepped) cw receiver such as ah

RFI meter, a spectrum analyzer, or an automatic network analyzer.

2. Feeding an rf burst through the antenna system and observing the response with

either a spectrum analyzer, an RFI meter, or a sampling oscilloscope.

3. Feeding the antenna system with a baseband step or impulse from any one of a variety of generators, such as a tunnel diode, a step recovery diode, a spark gap, or a mercury switch, and observing the response with a sampling oscilloscope.

4. Feeding broadband noise through the antenna system and observing the response

with a tuned receiver. Emission from radio stars has also been used as a standard

for calibrating large radio telescopes.

The above procedures will result in a measure of the antenna transmission or transfer

function, H(s), or the impulse response, h(t). The reflection coefficient, r(s), and the

analogous time domain parameter, y(t), the inverse Laplace transform of r(s), can also be

obtained. In terms of frequency domain scattering parameters, the transmission function is

given by S„^ and the reflection coefficient by S, ^

.
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Many studies have been made of broadband antennas. These have consisted of theoretical

analyses [1,2,3] and qualitative experimental measurements [2,3]. The parameters considered

have included boresight impulse response and transfer function [2,3], time domain antenna

patterns [2,3], antenna modeling with pole-zero singularities [1], and questions of reciprocity

[4.5].

The consensus in much of the research on antennas for impulsive field generation has

been that the so-called TEM horn is the most promising broadband antenna available. We have

developed our own version of the TEM horn which we feel has the best features of and some improvements over the earlier designs. Our main effort, however, has been a quantitative

evaluation of the field generation and measurement properties of the TEM horn which will

enable it to be used both as a standard impulsive field generator and as a standard receiver.

In this report we use the theory that is relevant to our measurement and add some modifications which we feel make our measurements more meaningful. We then describe the

measurement process that we have found most suitable to the development of a standard impulsive

field and give the design procedure and results of measurements on four balanced TEM horn

antennas. These antennas were constructed for use by the Army Redstone Arsenal at Huntsville, Alabama, and for further study at NBS . A resistive monopole developed here at NBS was used as part of the evaluation and is described herein.

2. FIELD MEASUREMENT EQUATIONS

We now derive two independent ways of rapidly measuring the boresight receiving characteristics of antennas like the TEM horn over a broad frequency range. These two methods are

compared and an error expression developed. The transmitting characteristics of the TEM

j horn is given. As a result, it is possible to use these horns either as sources of standard

fields or as standard receivers.

The first method of measuring the receiving function is illustrated schematically in

figure 1 together with the appropriate signal flow diagram where V,((jj) and V (cj) are the

input and output voltages, respectively, and E (cjj,r,0) is the electric field generated by

a conical transmission line at the location, r,9 of the horn. In this measurement the

resistive probe is not present. T and R^ are defined by the equations below, and the subf'pts

c and h denote properties of the cone and horn antennas respectively,

symbols T and R are used here instead of H that some other authors use since T and R are transfer functions in the usual dimensionless input-output sense. Henceforth, for

simplicity of notation, the functional dependence on to, r, and 9 will be understood but not explicitly written, and output input voltage ratios will be written as

E^(a),r,e) = T^(w,r,e) V.(a)) (1)

V^(u)) = R^(cj) E^(a3,r,e). (2)
Impulse

Generator

Horn 2

V.

1

OV^(co)

T^^(uj,T,d)

-o

R^^C.)

o

E „ (w,r,e) V (w ) n/ o

Figure 2. Measurement of T

h2'

Horn 3

V.

1

Horn 1 ^

13

Horn 3 Horn 2 ^

23

Figure 3. Measurement of R^ /R ,
S, = V (to) (from horn) /V. (to) (to cone)

he o

/ ^

X / \

when transmitting from the cone to the horn, for example. Therefore inserting eq. (1) and

the definition of S, in eq, (2) and solving for R, , \ = f \c- O)

c

Therefore, to measure the receiving function of the horn, one must measure the transfer

function of the cone horn combination S, and divide by the cone transmitting function T . he °

c The quantity T is defined by the usual expression for the field in a conical transmission

line [6] as outlined in section 4.1, Standard Cone Antenna, and the measurement of S, is

he

described in section 3, Measurement System. An independent way of measuring the horn receive function is obtained with two nominally

identical horns as shown in figure 2, with the appropriate signal flow diagram. This second

determination will be denoted R/^ . The subscripts hi and h2 refer to horns 1 and 2 respectively. The separation between the apertures of the two horns is r = r and for simplicity,

only on-axis gain will be calculated. For this arrangement

and

E = T V h2 h2 i

V = R^^ E,

(4)

(5) o Til h2"

Therefore, with eq. (4) in eq. (5) and solving for RJ' , one has

1 ^1 = T- ^12 (^^>

tiz

where S, „ is an abbreviation for S, ,, „. Solving for T, „

,

12 hlh2 ° h2

\2 = Rj^Sl2- <^^>

In appendix A it is shown that the far field, on-axis expression for T, „ is given by,

ikr ^ -,

.

o o

where n is the impedance of free space, Z is the characteristic impedance of the coaxial

waveguide that feeds the antenna, assumed to be real and frequency independent, (and also

277 across which V appears), A is the wavelength of the radiation and equals — times the

velocity of light, and k = 277/ A. This expression is equivalent in the time domain to the

transmitting function being the derivative of the receiving function [4,5]. The validity

of this relation was verified using a resistively loaded monopole which will be described

later.
Horn 1

Rt = S- /T

1 ice

Cone

Horn 2 Horn 1 r; = s^^iT^

Horn 3 Horn 2 ^2 ~ ^23^^3

Horn 3 Horn 1 ^1 = S13/T3

Figure 4 . Two independent measurements of R^
Therefore, substituting eq. (7) in eq. (6a)

K,

iZ Ar

o o K,

\2

^1

-ikr

^12 ^

(8)

The ratio,

"^ can be determined with a third antenna which we will take to be a horn

antenna for simplicity. The measurement setup is shown schematically in figure 3. In the

first measurement only horn 3 and 1 are present, and the voltage ratio, S-.„, is measured.

In the second measurement horn 1 is replaced by horn 2 in as closely the same position as possible, and the voltage ratio, S„^, is measured. Substituting these results in eq. (8)

and taking the square root yields,

ikr

^1 = iZ

o , ^13^12 — >^r -^

n o S23

1/2 -

e

(9)

This then is an independent determination of the horn receiving function and can be

compared with that obtained using the calculated field of the cone. The comparison is

summarized in figure 4. Having once determined R,

^

, one can determine R, „ from the ratio used in eq. (8) and

then using either eq. (6b) or eq. (7), one can determine T,„ and T . Having determined

these parameters, eqs. (1), (2), (4), and (5) can be used to determine the desired electric

field quantities. The above analysis was performed for TEM horn antennas, but it need not

be restricted to TEM horn antennas.

If desired, the emphasis of the above measurements could be rearranged to get independent

determinations of the cone transmission function, one determination would be theoretical

and the other would be experimental using parameters determined according to figure 4 with

R, determined as in the bottom half of figure 4. The uncertainty in the determination of R, , <5R, , by the two methods, can be found by

taking the derivative of eqs. (3) and (9) with the result that.

I

and

^1

'\l 6T 6S^

c he

ril c he

6S

13

6S

12

6S

23

13 12 '23 +ik

(10a)

(Sr (10b)

where errors in n> Z , and X have been assumed negligibly small. By careful definition of

the horn aperture plane, 6r may also be negligible. Additional sources of error may be

present due to near-field or off-axis effects which are not included in eq. (10), and will

be considered later. In any event, a conservative least upper bound on the measurement

error is the simple sum of all the individual voltage measurement errors, etc. However,

when determining the radiated field strength, the error in measuring the absolute voltage
< to

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level versus time and consequently versus frequency must be determined and added to the

above error bound. An estimate of these errors can be found in section 6.2 of this report,

but a complete measurement of these error limits will be the subject of a future report.

3. MEASUREMENT SYSTEM

The basic measurement system as shown in figure 5 is the same for each of the various

antenna combinations mentioned in the previous section. Figure 5 shows the transmitting

antenna being excited by a time domain waveform generator. The signal used can be a step

function, an impulse, or a variety of pulse shapes.

For calibrations on the ground plane, the antennas consist of a standard transmitting

cone with a resistive probe or TEM horn receiver. For measurements in a free-space field,

not associated with the ground plane, combinations of horns are used for both transmitting

and receiving.

The receiver is a sampling oscilloscope with a 50 ohm input impedance and a bandwidth

of dc to 18 GHz which was modified to interface with a minicomputer. The scope-computer

combination called the Automatic Pulse Measurement System (APMS) has been described by

Andrews and Cans [7,8,9]. The oscilloscope is a sampling device and requires that the

input be repetitive.

The operation of the oscilloscope is directly controlled by the minicomputer, except

for the time synchronization which comes from the generator. The computer determines the

sampling point, accepts the analog signal from the scope and digitizes it, moves the samling point to a new place on the waveform, and then arms the sweep circuits to be ready

for a trigger from the generator. The D/A and A/D converters in the computer are 12-bit

devices, and this is one factor that limits the resolution of the measurement and determines

the types of waveforms that can be measured

.

After the data has been stored in memory, the computer is free to manipulate it under

program control. One program presently available is the Fast Fourier Transform (FFT) which

converts the time domain data to frequency domain information. A time domain waveform is

measured by the system before it is transmitted, and the FFT is used to convert it to frequency domain information. A similar measurement is made on the signal after it is

transmitted and received again. The attenuation and phase delay of the transmitting and

receiving antennas together is then equal to the transformed received signal divided by the

transformed transmitted signal. This determines the quantity S, of eq . (3), for example.

Other programs are being developed to remove the effects of reflections which occur in

the environment around the antennas and to make the ground plane appear to be larger than

it actually is. One program makes use of homomorphic deconvolution to remove reflections

[10]; a second program is based on the Singularity Expansion Method, i.e., finding the

poles of an antenna system.

The various parts of the measurement system will now be discussed in greater detail.

Since the APMS presents some basic limitations on the type of waveforms it can measure, the

antennas and generator were designed to be compatible with the measuring system.

i^H
o-o

Remote

Sampling

Head

Sample

Time

Control

r- 1 Second

S&H

I ^ _ —/N

1

I "^

Averaging

Logic

I — 1

OSCILLOSCOPE

Difference

Amplifier

Control

^

Logic

J

D/A

Converter

71^

Interface

Logic

Minicomputer

A/D

Converter

Figure 6. Sampling scope - computer modifications,

10
3.1. Sampling Oscilloscope

The oscilloscope was modified specifically for antenna measurement use as shown in

figure 6. The solid blocks are original parts of the scope, dashed blocks are new parts or changes

.

The remote dual channel sampling head made it possible to sample directly at the

output port (or ports) of the receiving antenna without having large pieces of equipment in

the vicinity to distort the fields. The short duration signals are sampled and stretched

in the sampling head, and the narrow bandwidth signals which result are sent to the oscilloscope plug- in through 3 meters (10 ft) of cable. The time at which the sample is taken

is a function of the time of arrival of a trigger from the signal generator and a variable

delay in the oscilloscope controlled by the computer.

A second sample and hold stage is contained in the plug-in to the scope and usually

further stretches the input analog signal in time so that it can be used to vertically

locate a single dot on the CRT. In the unmodified form of the oscilloscope, each sample

produces one dot after another on the CRT face and the whole pattern is just this sequence

of dots with appropriate vertical deflections. In the modified unit, the oscilloscope

hardware can take repeated samples at the same time location of the waveform, and the

second sample and hold stage is used to sum up all these inputs. Averages of 1, 10, or 100 are possible.

After the correct average is taken at one location of the input waveshape, the computer

can either change the delay time of the sampler to measure a new part of the input waveshape

or request a repeat of the previous measurement for digital averaging. Since the input

signals are quite small, this averaging can produce the improvement in signal to noise ratio

required for typical antenna measurements. The noise reduction permits the signal to be

displayed directly on the CRT without having to be processed by the computer. This allows

the operator to find received signals that might be buried in the noise in real time. The

analog signal is then processed by the computer's A/D converter, and the information is

stored in the core memory of the computer. The maximum rate of the computer's A/D conversions is about 25 kHz

.

Each scope channel is averaged separately and then can be combined as A + B or A - B before being digitized. When antennas are measured on the ground plane, signals are measured

with respect to ground using only one 50 ohm coaxial input, either channel A or B. Antennas

designed for free-space field measurements, that is measurements that are not referenced to

a ground plane, are balanced, and the two output signals are 180 degrees out of phase. These

are measured by applying one to channel A, the other to channel B, and measuring the

difference in the outputs of the two channels (A-B) . This presupposes that the two channels

are matched in gain and phase which is the case at the frequencies being used.

11
15V _

Time Jitter Error

Q Quantized Data

Actual Waveshape

'' Errors Due to Jitter

o

> lOV-

>

iJM

C

a

3

c-

(U60

ca

5V--

CD 6

I 4 u

so

M

Ou

rt „

jj 2

cu o

-2

Quantizing Error

OV f

^—I—

^'—

(^ <ti P—#

<|i (t.(t(t> (l)<t(t> (ti O0Od)<> |t

>

Time Increments (seconds)

Figure 7. Quantizing and jitter errors.

==

^^1

1 /^^^^^^^-^-^^-^ -^---^^^^^-^

2 3 4 5 Time (nsecs)

Figure 8. Driving impulse to the transmitting antenna.

12 3
3.2 Computer, Peripherals, and Analog Converters

The minicomputer used in the system was slightly modified to interface with the

oscilloscope logic as mentioned in [7]. It was a commercially available unit with internal

12-bit A/D and D/A converters.

The peripherals needed with the computer are shown in figure 5. The terminal has a graphics capability, and a fast hard copy device is also necessary. Most programs and data

files used during a measurement are large and the additional memory of a floppy disc is

needed to handle this volume. A detailed description of the peripherals will not be given

here since they do not directly effect the accuracy of the measurements, assuming that they

function properly.

Since the sample time position is controlled by the computer's D/A converter, the

number of samples taken in the measurement time window is a function of the number of discrete

voltage levels which the converter can produce. A 10-bit converter has 1024 discrete

levels and could produce that many different delays, each sample point being separated from

the next by only one bit of information. If the accuracy of the converter were ± 1/2 bit,

the separation between adjacent samples might vary from zero to twice the intended size,

giving rise to rather large errors. For this reason, a 1024 point sample sequence is

better produced by a 12-bit or preferably 14-bit D/A converter. The 12-bit converter

available in the minicomputer provides a sample density of 1024 points per trace with an . adequate accuracy, but this becomes the major limitation on the type of waveforms that can

be analyzed, since the error in time can cause a large error in amplitude when the slope of

the waveform is large.

In order to make a correct measurement, the entire waveform must be sampled, that is,

it must fit in the measurement window. Consider a waveshape composed of a fast transition

followed much later by a significant reflection. In order to record both the initial

response and the reflection in the same time window, the sweep speed may have to be slowed,

allowing the possiblity that the transition itself is too fast for the sampler to generate

an adequate number of samples during the transition time. For antennas having this characteristic, error in the measurement can be large.

A second type of measurement error occurs if the waveform has a component with a very

small amplitude but lasting for a long time. The energy is appreciable and will contribute

significantly to the spectrum but may go undetected by the sampler because its amplitude at any one point in time is not large enough to move the A/D converter from one quantizing

level to the next. Some impulse generators produce a waveform like that of figure 7 . If the impulse is

15 volts high and 2 samples wide, the area under the impulse curve equals

y X 15 X 2 = 15 volt seconds.

The impulse is followed by a negative undershoot that has an area equal to 1/3 the area under the impulse but has a small amplitude. It is possible that the digitized data would

miss the undershoot entirely.

13
300

240

180

120

> 60

4J3

3O

-60

CO -120

0) ^

-180

-240

-300

1

A jiA A A - ]

f\ V>—^mvvy>y>./^-'

I

I 12 3 4 5 6 Time (nsecs)

Figure 9. Output voltage from receiving antenna.

*! K 9=4°

Horn

Under Test

Figure 10. Standard field from cone antenna.

14
With 12-bit A/D and D/A converters, the vertical and horizontal scales are divided

into 4096 discrete digitizing levels. As a rule of thumb, in this case, the waveshape

should occupy more than half the vertical space of the CRT and should be completely finished

in a time not to exceed 300 times the duration of the fastest transition to be recorded.

It is desirable that the undershoot end within a few pulse durations. The impulse waveform

used in our measurements is shown in figure 8 and can be seen to be finished in a time less

than 25 times the risetime. The received output of a pair of TEM horns built for the Army

Redstone Arsenal is shown in figure 9 and has an internal reflection which occurs 3 ns after the main response. The total response is finished in about 80 times the risetime.

Both the main response and the reflection are easily measured by the APMS . An additional

refection occurs as an interaction between the horn and balun which occurs 12 ns after the

major response but is not recorded in the time window. This could be measured on the

APMS except, for this large a time, other reflections within the measurement environment

interfere.

4. ANTENNA SYSTEMS

4.1. Standard Cone Antenna

The conical antenna was chosen as a standard to provide a reference measurement system

that is better defined theoretically than the TEM horn, and therefore whose accuracy can be

determined more completely. Figure 10 is a schematic of the cone located over a ground

plane, and figure 11 is a photograph of the 270 cm (8.9 ft) long cone over an early version

of the ground plane and showing the APMS in the background.

The cone is driven from a generator with a 50 ohm resistive source impedance. The

impulsive signal is brought up from under the ground plane on a 3 mm semi-rigid 50 Q coax

line and propagates out along the cone forming a spherically expanding wave of short

duration. For a length of time less than the propagation time along the cone, the conical

antenna presents a constant resistive impedance to the expanding electromagnetic wave

throughout its entire length and looks exactly like a biconical transmission line. In effect, the energy propagating from the 50 ohm line simply sees a resistive mismatch at the

base of the cone, and the two form a frequency insensitive voltage divider without introducing waveform distortion to the time domain waveform. The field can then be described as an attenuated replica of the driving voltage.

The impedance of the cone over a ground plane is a function only of the cone half

angle (6 ) which is chosen at 4° for ease of fabrication. Smaller angles produce too long

and thin a point while larger angles produce too large a disc at the top end. The characteristic impedance is given by [6]

Z = 60 In cot (e /2).

^'"'"^ o o

15
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OJ

4-1 c

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If 6 =4 degrees, the characteristic impedance (Z ) becomes approximately 200 ohms,

o ° Figure 12 shows a TDR measurement of the cone indicating that its impedance is close to

200 ohms.

Knowing the impedance of the cone and the fact that it is resistive, allows a simple

calculation of the voltage applied to the cone. A direct measurement of the output from an impulse generator is usually made with an oscilloscope having a 50 fl input impedance and

yields

V(t) (R + 50)

V (t)

gen

meas. gen

50

(12)

The voltage appearing on the cone with respect to the ground plane is

V (t) 200

V (t) = -S^3

cone^ '^ R + 200

gen

Substituting eq. (12) into (13) gives

4V(t) (R + 50) V (t^ = ^eas gen

cone^ '^ R + 200

gen

(13)

(14)

Special care has been taken to ensure that the NBS built generators have a 50 ohm impedance

over the frequency range of interest, and in this case eq. (14) reduces simply to

V(t) = 1.60 V(t) cone meas

.

(15)

Given the voltage between cone and ground plane as a function of time, the field in the

TEM mode at a point in space can also be given as a function of time. From [6] for a biconical transmission line in the frequency domain,

rE„(cu) = —n sin 6 ^^i(a,t-kr) ^

3^i(a.t+kr)

(16)

and the voltage between the pair of cones is given as

C

( V(a)) = 2n In cot - ^^i(a3t-kr) ^

g^Ko^t+kr) (17)

Substituting (17) into (16) and V (co) = V(w)/2(over a ground plane),

E,(w) = V (co) cone

r sin (e) In cot

or in the time domain

(18)

E^(r,e,t) = V (t) cone

r(sin 9) In

17

cot

(19)
200

50-

204 Si 170 ?.

Second Reflection Due to

Itiput Connector

Mismatch at Input Connector

-H 1—

200 AOO

Distance (cm)

600

Figure 12. TDR trace of conical antenna.

U- 36 <m >|

3 mm Semi-rigid Coax

Coax Center

Conduc ter Soldered to Copper

Styrene Foam

"'5^

Ground

Plane Figure 13. Half horn model .on ground plane.

18
where,

E.(r,e,t) is the E field as a function of time in the 9 direction and at a distance r

6 from the apex of the cone,

V (t) is the voltage between a reference point on the cone and the ground plane

cone

^ ° measured along the E field,

r is the radial distance from the apex of the cone at the ground plane to the point

in space at which the field is desired,

9 is the angle between the axis of the cone and the radius r,

9 is the half angle of the cone. The most important observation to make at this point is that the E field has exactly the same shape in time as the driving voltage from the impulse generator. In other words, the cone produces a field which is a high fidelity replica in time of the driving voltage. A high fidelity receiving antenna would then produce an output voltage time waveform which

nearly resembles the generator time waveform, and this is used as a test for the quality of

the antenna.

4.2 TEM Horn Antenna

The antennas chosen for the impulsive free-space field measurements (not on the ground

plane) were the TEM horns as mentioned previously. Besides being mechanically rugged, they are quite broadband in the receiving mode, both in magnitude and phase.

A half-horn was constructed as shown in figure 13. The antenna was made by cutting

the proper triangular shape into the copper foil on a standard 1/16" glass epoxy printed

circuit board. Holes drilled on the edge of the board are used to fasten the antenna to the ground plane with 4-40 nylon bolts, and the wedge spacer was cut from polystyrene foam.

The coax is again the 3 mm semi-rigid coax cable with a type N connector.

Figure 14 shows the time domain receiving response of the horn when the cone was used

for transmitting. The similarity between transmitted (Fig. 8) and received impulses attests

to the good fidelity of the horn as a receiver. The frequency domain response shown in

figure 15 confirms that the response is reasonably flat (± 3 dB) from 200 MHz to 4 GHz. On

the basis of this type of data, the TEM horn was chosen for use in free-space field measurements.

It is shown in the appendix that for any antenna the far-field transmitting response

is exactly a constant times the derivative with respect to time of the receiving response.

In other words, in the transmit mode, the low frequency rolloff with decreasing frequency

is exactly 6 dB/octave steeper than in the receive mode. In order to verify these results,

a resistive probe (monopole) with a flat receiving response was built to be used as a standard receiving antenna. The TEM horn was then tested in the transmitting mode with the

resistive probe as receiver. Details of the probe construction are given in the next

section. Figure 16 shows the probe output waveshape when the horn was transmitting. The

doublet resembles a time derivative of the impulse driving the horn. The frequency response

of the pair, figure 17, shows a rapid rolloff toward the lower frequencies. It was also

19
3a

3O

C

c

0)

4-1

-1

Time (nsec)

Figure 14. Receive time domain response of horn model.

-32

-34

OJ -36

>

,o

ca -38

PQ•O

N—

-40

(U

tnC

o -k7

a.

en

(1) Pi -44

(1) >

•H -46

0) pi

4

A^/I Ax Wl 1 M^\j .A...

V\ A

vl

2 3 4 Frequency (GHz)

Figure 15. Receive frequency response of horn model,

20
16

%

3

a.a

o

n) -8

c

0)

-If

-24

/I

/\

A A a ^:i

V

' -'»•' ^ V "^

'

V

2 4 Time (nsec)

Figure 16. Time domain response of horn-resistive monopole pair.

+10

c (U o >

p. O

to X>

(\) ca Pi m

4-1 •o e

U)C

nj

)-i H

-10

-20

-30

-40

->.Aa \

v^

y^^

ry

r

Jf

1

2 3 4 Frequency (GHz)

Figure 17. Frequency domain response of horn-resistive monopole pair,

21
N-

-A ^

~

-~^

i

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/ / / 1/ A / / /

/ / //1 / \./ / /

/ / / /

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/

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/ / /

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/ / / / / / y /A

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k

<-

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01 X Q.

nJ D. o O u u ^ Uh o

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4-i ex, tj < 0) 3 r-1 X) o 00 c U c

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tfl cd •H c

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23
shown empirically that the transmitting function of the probe was the derivative of its

receiving function and the receiving function of the conical antenna was Indeed the integral

of its transmitting function.

TEM horns for free-space field measurements were made as nearly Identical as possible to the half-horn ground plane model. A small ground plane was made a part of the horn structure

to aid in making TDR impedance measurements of the horn after construction. The structure

is shown schematically in figure 18. The ground plane is longer than the antenna, and has

the same width as the antenna printed circuit boards. The actual antenna is shown in the

photo, figure 19. Each half of the horn is brought out separately on its own 50 ohm coax connector.

This type of structure makes it easier to test each half-horn separately on a 50 ohm TDR

and makes the antenna more nearly the same as the half -horn tested on the ground plane.

Ferrlte beads around the coax and an absorbing foam are used to damp the reflections on the ground plane that propagate onto the outer surface of the coax lines.

The antennas can then be driven through a balun which splits a single unbalanced input

into two identical outputs of opposite polarity. A better way to drive the antenna would

be from a balanced generator that produced the two signals directly. In the receiving mode,

the two outputs can be combined in a balun. Again, the better way is to look at the two outputs directly on two channels of the sampling oscilloscope (A and B) , and since they are

of opposite polarity, they can be combined as A-B. The balun is a simple broadband power divider with one output delayed and Inverted

while the other is simply delayed, see figure 20.

Each arm of the divider Is 30 cm long, and the shields are grounded together close to the connectors. The resulting inversion joint is shown in the photograph on figure 21. The whole balun is mounted in a 7 cm x 10.5 cm x 16 cm aluminum utility box. The extra

space in the box is filled with absorbing foam to reduce the effect of reflections. The

time domain response is shown in figure 22 with the non-inverted output from the balun

inverted in the scope for comparison with the inverted pulse. The slightly smaller waveshape is due to the loss at the inverting splice and will have a slight degrading effect

on the balance of the horn. Figure 23 shows the frequency response of a pair of Identical

baluns connected together in such a way that the inverted signal from one balun passes

through the second without additional inversion and vice versa. In figure 24, the frequency

was swept from 10 to 1200 MHz and the resultant variations are less than ± 3 dB. The

ripple is caused by multiple reflections between the two baluns but the average frequency

response is flat over this frequency range.

Free-space field testing of the TEM horns was done in two steps. The receiving characteristics were measured by placing the horn in the spherical field of the conical antenna

but above the ground plane as shown in figure 1 and 25. Then the transfer function of a pair of horns was measured on a small laboratory range, see figure 26, where the antennas

are separated by 2.5 meters and are positioned 80 cm above the floor. Results of these two measurements could then be used to determine the transmit function, T , and receive function,

R, , as explained in section 2.

24
s.^ c

0)w3aeo3

•H

25

lu.
o

•H

4JO

S-i 4-1 WCoo

q!

=) m

•H

26
+1

en

3

-1 -

& -2 -

-3

-4

200

Time (psecs)

400 600 800 1000 1200

iTipiit Impulse 90 psi-c . Duration

+ Noninvertcd Output of Balun Which

Was Inverted in Scope For Comparison

- Inverted Output of Balun

Figure 22. Comparison of the time domain responses of the balun.

PQ

to 5 12

16

li'ViLmI

1 VMn

1

V

2.5 5.0

Frequency (GHz)

7.5

Figure 23. Frequency domain response of baluns.

Figure 24. Frequency domain response of baluns using

swept frequency techniques.

27
^'^"^^ ."*^-.^*

.f ^ > ^'^'^-asT^j!

Figure 25. Calibrating a horn over the ground plane.

28
I

Figure 26. Measurements on the TEM horn.

29

»^mmi^mmlm
0)aH o

±.

H 2h- T (J

en 0) en Xi

.H

H

O eou

•H o&ocoe

-a

(U -a

n)oH

<U>

•H

10

0) P5

CO

0)

»^3

60

•Hc3O

60

0) 42oaocoB

0)>

•rH

•rlCC

(U

erf3

60

30
I

For the two horn measurement, the most serious reflection comes from the floor surface.

Since the path length from one horn to the other by way of a single reflection from the

floor is three meters (a half meter longer than the direct path) , reflections from the

floor arrived at the receiver only 1.5 ns after the desired pulse and could interfere with

a measurement of the transfer function. The floor was covered with microwave absorber as

in the photo which reduced the effect of the reflections.

4.3 Resistively Loaded Monopole

The wave generated by the conical antenna is spherical, and an accurate measurement

will be done best if the receiving antenna is small in all dimensions. Considering a dipole

(monopole on the ground plane), however, the greatest sensitivity is obtained near the

resonant frequency which is a function of its length. Very small dipoles driving a low

impedance, say 50 ohms, can be expe::ted to work best at frequencies near resonance and not

be particularly broadband. This should not be confused with statements indicating that

capacitively loaded miniature dipoles are broadband. Small metal dipoles are broadband if

the only load they must drive is a small capacitance. These are useful for measuring cw or peak pulse fields, but this broadband quality disappears when the dipole is resistively

loaded by a 50 ohm measuring system. The impedance of such a dipole is capacitive and

becomes very high at low frequencies making a poor match to 50 ohms.

A resistive monopole was developed at NBS which is small in size and yet broadband in

a 50 ohm measurement system. It is described by Kanda [11] and its use in the conical

transmission line is shown schematically in figure 1 and in the photograph of figure 27. It mounts through the ground plane with a type N coaxial connector bolted to the under

surface. All connections can be kept under the ground plane.

The monopole is constructed of a 2.3 mm diameter glass tube as shown in figure 28. The lower half centimeter has a gold band for making connection to the center pin of the

connector. Gold epoxy works well to make the connection. The next 7.5 cm is vacuum coated

with Nichrome in such a way that the resistance nearer to the connector is about 30 ohms

per cm. Toward the free end, the resistance increases to about 200 ohms/cm (a higher value

would be even better) . The tapered resistance was obtained by making the deposition with

the low resistance end of the substrate much closer to the metal source than the high

resistance end. In its present design, the monopole is too fragile for field use and is

limited to laboratory applications.

Figure 29 shows a plot of the transmitted impulse and that received by the resistive

probe. The similarity indicates that the probe is a high fidelity receiver, and this is

further borne out by the broad response indicated in the frequency domain, figure 30. The

response of the probe to a plane E field was obtained numerically by Dr. Kanda and is shown

in figure 31 together with the experimental results. The agreement was considered adequate

for use as a standard receiver on the ground plane. Other antenna types can now be measured, (1) as a receiver using the cone for a standard transmitter, and (2) as a transmitter using the resistive probe as a receiver. The antenna of principal interest is, of course, the TEM horn.

31
0)c

ou

u 3 3

a. 5 2

%

CD rHO

o

-• C^ 2 e

o

aj-i ° -1

-2

-3

Figure 29. Time domain response of resistive probe.

32
P313

(U

COc

o

a.

a)

-49

-52

-55

-58

-61

-64

-67

1,25 2.50 3.75 5.00

Frequency (GHz)

6.25

Figure 30. Transfer function of cone-resistive monopole.

-20

-30

-40

-50

-60

-70

<: -80

I 1 1 1 1

^ X*

-

' •

- X MOMENT METHOD

• WU-KING O EXPERIMENT r

1 1 1 lOOkHz IMHz lOMHz lOOMHz FREQUENCY

IGHz lOGHz

Figure 31. Calculated and measured response of resistively

loaded monopole.

33

m
5 . GENERATORS

As mentioned previously, many different types of signals can be transmitted by a broadband antenna system. To test the impulse response of an antenna pair, the appropriate

driving signal is, of course, an impulse. The major part of the work on this project has

been done using an impulse generator specifically designed by Dr. James R. Andrews. The

design was described at the 1975 IEEE Symposium on EMC and is contained in the proceedings

of that symposium (Oct. 1975) [12]. Dr. Andrews has included all the design information in an NBS Report written with M. G. Arthur, titled "Spectrum Amplitude, Definition, Generation

and Measurement" [13].

The generator output has already been shown in figure 8. The impulse is approximately

10 volts high and 180 ps wide at its base. The spectrum is shown graphed in figure 32 and

tabulated in figure 33, and can be seen to be flat within ± 4 dB from 100 MHz to 4 GHz, and

is only 14 dB down at 7 GHz. The generator impedance is nearly 50 ohms due to the backmatching effect of a directional coupler used in the output stage. Since the horns being

driven by the generator have a substantial reflection at the radiating aperture, it is

necessary to maintain this backmatching to reduce multiple reflections.

To extend the work to higher frequencies, it is possible to amplify the impulse in a Traveling Wave Tube Amplifier (TWTA) . The output waveshape no longer resembles an impulse

but still has a relatively broad spectrum. Using a C band amplifier, one can generate usable

frequency components from below 2 GHz to well above 12 GHz. The output spectrum of an X band amplifier covers 2-16 GHz with only one serious resonance at 9 GHz. For more

information, see Dr. Andrew's report [13] mentioned previously.

It has also been mentioned that the TEM horn differentiates the driving signal when

it radiates. If the measurement calls for a broadband impulsive EM field (for example,

to test the impulse response of a receiving antenna alone), it would be better to drive

the transmitting horn with a fast rising step function. A long time duration square wave

having extremely fast rise and fall times will give equivalent results. Figures 34 and 35

show a square wave and impulsive transmitted wave forms, respectively. The differentiation

produces a field composed of a series of impulses having alternating polarity. Experiments

have been done with square wave sources with favorable results, and are reported in the next

section.

34
71

p. >

01>o

.a

cd m

T3

64

57

50

enPo 43

o 36

pC/3 29

'

' f — -

i.

5.0

Frequency (GHz)

10.0

Figure 32. Spectrum of Andrew's impulse generator.

SP

*WA;/EF0RM TirPET -1=STEP. O^IMPULSEe

WAltEFCKtl ACQUISITION MODE? 1=SWEEP SEQUENTIAL*

0=POINT SEQUENTIAL!

N'JMBER OF DUMMY DATA ACQUISITION SWEEPS »?30

NLitieER OF UAUEFORM POINTS =?512

NUMBER OF TIME-A<,€RAGED WAVEFORMS*? 108

UERTICAL SCrt-E FACTOR IN rHVCM=?20ee

TIME SCALE FACTOR IN NSEC^CM =?1

INDIUIDUAL SPAMP LISTINGS? (l='r^3> 0=NO)1

NUMBER OF FREQUE><CY DOMAIN AUERAGES=?1

y-AXIS CALIBRATION-? <1=YES, 0=NO)0

X-AXIS CALIBRATION? <1=YES, 0=NO)0

LISTING UNITS? -1*06, 9=U-PS -1 ^ 108.000 WZ 60.212 Oe

200.000 MHZ 60.641 06

308.000 MHZ 61.940 06

400.000 ^fC 63.511 OB

500.000 MHZ 65.006 oe 600.000 MHZ 65.882 06

700.000 IfC 66.288 DB

800.000 mz 66.676 06

900.000 MHZ 66.656 OB

1000.000 MHZ 66.449 08

1100.000 MHZ 65.826 06

1200.000 MHZ 65.448 OB

1300,000 MHZ 65.964 DB

1480.000 MHZ 66.351 DB

1500.000 mz 66.281 DB

1 600.000 MHZ 65.880 DB

1700.000 MHZ 66.869 DB

1800.000 MHZ 66.433 DB

1900.000 MHZ 65.945 DB

Figure 33. Spectrum of Andrew's imp

35
500

400

300

200

100

00C

•H

•H

I -100

nJuH

n -200

^ -300

-400

-500

-^

fl^'—

^

'

'--^ '

40 80

Time (nsecs)

120

Figure 34. Square wave driving transmitting antenna.

>

e

500

400

300

cfl 200

cc

td 100

>g -100

Pi

M-. -200

o

g. -300 ° -400

-500

rt \ i ^ Ia

If If iTW /Tl |U

n

'

I

1

40 80

Time (nsecs)

120

Figure 35. Signal received from a transmitted square wave.

36
6. MEASUREMENT RESULTS

6.1 Description of Measurements

Before the radiating characteristics of the horn antennas could be measured, the

standard cone had to be evaluated. The impedance was measured using the APMS, and the

magnitude lies between the limits of 140 and 270 ohms, and errors from this source are discussed in the next section. The 30 cm length of coax-feed line to the cone antenna has

a loss which is a function of frequency. This was also measured with the APMS, and the

average loss is 0.3 dB with a maximum deviation of ± .3 dB. By assuming that the antenna

impedance is a flat 200 ohms and correcting the coax loss at a flat 0.3 dB, an error of

approximately ± 1 dB can accumulate.

Now the cone transmit function is calculated for the E field at any point, P(9,r), on

or above the ground plane. For the layout shown in figures 10 and 25, the distance r is

2.5 meters (adequate for far field). With the horn antenna positioned at 16° above the

ground plane, 9 will be 74°. For a 200 ohm cone impedance, the voltage on the cone is 1.6

times the incident driving voltage. Therefore, the transmit function can be calculated by

substituting into eq. (18) and using eq. (4),

" V„„„„('^) r Qq] 2.5 (sin 74.2°) In (cot 2°) V cot -j\ meas . ... , r sin(G) In

Measurements made over the ground plane as shown in figure 1, include the combined

transfer function of the cone transmitting and horn receiving functions or T R, . When this

is expressed in dB, the product becomes the complex sum T ,„ + R, ,„ where T ,„ = 20 Loe^„ T , cdB ndB cdB 10 c

R, , is similarly defined, and these are complex Logarithms. However, our calculations of

R, , , T , and T, will only include the magnitude and not the angle. In dB, the cone transfer function becomes -14.1 dB referenced to 1 —tj—- Reducing this by the 0.3 dB coax

loss yields

^cdB

= -1^-^ dB

^ . (20)

To calculate the receiving characteristic of the horn alone, we simply subtract the cone function or

\dB = f^cdB -^ ^dB^ - ^cdB = f^cdB ^ ^dB^ ^ ^^'^ ''^

As mentioned in section 2, there is another method for determining the receive

function using three horn antennas. Measurements were made using several combinations of

the four available horns. Substituting in eq. (9), the values of S,^ and S,„ for S^ „ and

S2„, respectively, (using horn 4 instead of 3)

37
T)o

Xi

4-1

(1) s

-vo 0) SI ^

•u o

a) u

2

13

Cvi )-i C ffl C t:

QJ c U m

C a-i < CO

o

c

D

m o o

HI (-/A I 3Aoqv gp) Bsuodsay

oH

o

CO

(3o

•H

4-1 O

C

3

(U

on

c

(UM

(U

M-l

o

c

o

to •HM

tfl

i-Oo

00

38
K

I _ iZ S,^S^„

o , 41 12 Ar

n o S^2

1 /o ikr

1/2 o

e

^ . (21)

The magnitude of this receive function has been calculated and is shown together with the

first method for horn 1 in figure 36. The difference is probably due to near-field effects,

the cone transmitting function, and reflections, to mention just a few sources. Data

determined by these two methods were averaged and are shown for horn 1 and horn 3 in

figures 37 and 38, respectively.

Measurements of the transmit function according to the method in eq. (6b) are made by

transmitting from one horn antenna and receiving on a similar antenna. The transfer function

is again a combination of T, R, , and expressed in dB, these become T, , + Ri jt,- To determine

the transmit function alone, we simply subtract the horn receiving function determined in

the previous experiment or

hdB f^dB "'' ^dB-' \dB' (22)

Now since there are two determinations of the receiving function, R^ measured with the

standard cone and R! measured by the three antenna method, there can be two different

determinations of T, . This method is really a deconvolution of a receiving function from

a combination of transmit and receive functions. Accordingly, the two determinations will

be distinguished by calling the first cone deconvolution, and the second three antenna

deconvolution.

As mentioned earlier in this report, a second method is also available for determining

the transmitting function of the horn antenna. It is shown in section 2, eq. (7), that the

transmit and receive functions are related by the following equation,

nR^

Z Ar

o

For example, at a frequency of 4 GHz and at the normalizing distance of 1 meter, and a 50 ohm source, |t, | = 100.5|r, |. In dB, this becomes

\dB= \dB + ^0-°^» (23)

In this method the transmit function is determined directly from the receive function

and it will be referred to as the direct method.

Results of this determination and the two deconvolution determinations for horn 1 are shown in figure 39 for comparison. Note that the two deconvolution curves agree to about the same degree as the receiving functions do. The direct method data disagrees

more, especially at the higher frequencies as expected due to the increasing uncertainty in

the standard cone transmit function at those frequencies. The three determinations were

averaged for the final data and these are shown for horn 1 and horn 3 in figures 40 and 41,

respectively.

39
>

Hi>o

•a

w

co

ft

0)

-30 -1

-35

-40

-45

-50

-55

\^-

2 3 4 Frequency (GHz)

Figure 37. Receiving frequency response of horn 1.

>| e

>

o

(Uto

p:o

ft

CO

(U Pi

-30 1

-35

-40

-45

-50

-55

3 4 Frequency (GHz)

Figure 38. Receiving frequency response of horn 3.

40
« • < co

•H

C

4-J o c 3 •H o

rH 4J •H O 3 M

l> iH

U

C o 0) O > > O c c

(U o o n a

0)

u • P 4J U o ^

0)o •H

0-) o O couo

CO

0)

ca

flo

p.

COu od

Qj3

cr

(1) P4 S

(UMm

•H

r+ o

CM

(A/—I SAOqv aP) 3Suods3-a A o

CO •H

cfl P.Soo

(U

V400

•H

41
+5

>l E

0)>

o<

PQT3

o

CO

Pi

>

0)o

COc

o

p.

CO Pi

-5

-10

-15-

-20.

Averaged Data

Estimated Error Limit

Frequency (GHz)

Figure 40. Transmitting frequency response of horn 1.

+5

-5-

-10- Averaged Data

Estimated Error Limit

-15-

-20-

3 4 Frequency (GHz)

Figure 41. Transmitting frequency response of horn 3.

42
6.2 Evaluation of Errors

As shown in eqs. (10a) and (10b), there are different sources of error for the two methods of measuring R. This is also true for T. Consider first, the determination of a horn receiving function from measurements with the standard cone. Equation (10a) divides

the sources of error into those associated with the cone transmit function, and those

produced during the measurement of the cone-horn transfer function S, . The cone transmit function is determined theoretically, and is well documented. The one major source of error, as was stated previously, is the impedance of the cone and its

input transmission line. These effect the voltage dividing ratio directly, and when multiple

reflections between the generator and the base of the cone together with the error in

measuring the impedance are included, there is an uncertainty of ± 0.7 dB in the total

transmission function.

There is an additional error caused by losses in the input transmission line and these

losses together with the uncertainty in their measurement are estimated to be an additional

+ .3 dB. A large part of these errors can be measured and calibrated out. This will be

done for future work, but for the present time, the entire effect is included as an uncertainty to give a total of ± 1.0 dB. In addition, errors in the measurement of S are

caused by the limitations of the APMS and have been estimated to be approximately ± 1.0 dB over the frequency range of these measurements [12]. The total uncertainty is ± 2 dB when

the antennas are used in a laboratory environment, that is, where both the external (room)

reflections and the internal ones due to the balun have been eliminated or time windowed

out. This is the way we have normally used them.

When the antennas are used in the field with some type of EMC receiver the balun is

necessary and its reflection cannot be time windowed. The reflection occurs in the

following way. A signal propagates from the driving point of the antenna and reaches

the aperture, where it sees an impedance higher than 50 ohms and is reflected back to the

generator. However, as it enters the balun, there is only 6 dB isolation between the two antenna ports, and some of the reflected signal couples to the other side of the antenna

and repeats the reflection. Since it requires 12 ns to complete this "reflection cycle"

and the maximum duration time window presently obtainable in our facility is only 10 ns, the effect cannot be directly measured. An estimate has been computed from the measured

reflection coefficients of both the antenna and balun. The most serious effect is produced

at lower frequencies because the antenna is less efficient and its reflection coefficient is

higher. Since the error is changing rapidly with frequency, an accurate estimate is very

difficult, and for this reason it is not calibrated out. An uncertainty of ± 3 dB is incurred by this effect below 1 GHz and ± 2 dB above 1 GHz. This is then added to the other two errors for a worst case total of ± 5 dB below 1 GHz and ± 4 dB above.

Now, consider the determination of the receiving function by the three antenna method.

Equation (10b) shows that the sources of error will be the measurement of the horn transfer

43
10 cm Separation

AO cm Separation

160 cm Separation

erf u

10 r

15 -

20 -

^ 25

30

35

40

45

A/'v\ >

50

.2 1.0 2.0 4.0 6.0

Frequency (GHz)

Figure 42. Effect of antenna separation on frequency response.

44
I

functions S^-, S^„, S„. and the measurement of the distance r . This last factor can be 13 12 23 o measured accurately and the error considered negligible. Each of the transfer functions

have a potential error due to the ± 1 dB uncertainty in the APMS. Equation (10b) shows

these three errors and indicates that the S„ error subtracts from the other two and this is

indeed the case for systematic errors where the measurement system introduces a bias of the same sign to each transfer function. However, the random error in each transfer function

does not necessarily have the same sign at all times and on the average does not subtract.

Simply adding the three errors together for a worst case combination gives ± 3 dB. This

will be the total error under the laboratory conditions discussed above.

For field measurements, the balun reflection error must again be included. The total

uncertainty in the measurement increases to ± 6 dB below 1 GHz and ± 5 dB above.

Figure 36 shows that the receiving function of horn 1 determined by these two methods

not including the reflection error, has a maximum difference over the frequency range

600 MHz to 5.5 GHz of ± 2 dB, which is within the estimated error bounds of each system

individually

.

Finally, an error is introduced when the receiving antenna is not in the far field

of the transmitting antenna. These effects are estimated by measuring the transfer function

of a pair of horns at several spacings (see fig. 42). In changing the spacing of the antennas,

the pattern of reflections from the walls and floor also changes, so that not all the variations in transfer function can be attributed to near-field effects. However, since the

data taken with a 10 cm spacing between antennas diverges from the 40 cm and 160 cm data

above 2 GHz, it can be safely said that the near-field effects over the frequency range of

600 MHz to 5 GHz exceed ± 3 dB when the spacing is reduced to less than 40 cm. The larger

error occurs at the higher frequencies where the near-field region extends farther in front

of the antenna than at lower frequencies. At distances of 2.5 meters, at which our measurements were made, the near-field effect should be small (at worst .1 dB for cone transmitter).

The transmit functions for the horns determined in the two ways each have their own particular sources of error. In the first case, the uncertainty in [T, , + ^uj,,] is given

by the uncertainty in S , . The uncertainty in R, ,„ is described above. The total uncertainty for this determination is estimated to be the sum of the two or ± 3 dB over the

frequency range.

A third determination of the transmit function is calculated from the receive

function according to Kerns [16] and has exactly the same uncertainty as that function.

Therefore, the estimated uncertainty at worst is again given as ± 3 dB over the frequency

range. In both cases, however, if balun reflection errors must be included, then the total

worst case uncertainty increases to + 6 dB below 1 GHz and ± 5 dB above 1 GHz. Notice that the measured value of the received function appears in eq. (23) with a positive sign, while in eq. (22) the sign is negative. It should be expected then, that any

systematic errors in measuring R^ will have the opposite effect in the two determinations

of T, . The three curves shown in figure 39 agree well in the range of 600 MHz to 2 GHz, to within 2 dB up to 4 GHz, and no worse than 3 dB to 5.5 GHz. The increasing difference is

45
attributed partly to the small near-field effects which become more pronounced at high

frequencies, and partly due to errors in the cone transmitting function. Notice that the

determination based on the three antenna receive function lies between the other two curves. The fact that the three antenna method does not require the standard cone indicates that

this difference is probably due to the cone. These results verify the measurement technique

and give an estimate of the error.

7. RECOMMENDED USE OF ANTENNAS FOR FIELD GENERATION

The horn antennas were designed to permit the user to establish a known impulsive field

on his own antenna range. The horn selected as transmitter is positioned on the range

between 2 and 3 meters from the device being evaluated (usually an unknown antenna) . The

antenna is driven from a well-matched 50 ohm waveform generator, either impulse-like

or sinusoidal, through some length of 50 ohm coaxial transmission line. The spectral

intensity (or output voltage for sinusoidal generators) must be known for the generator and

cable into a 50 ohm load.

The unknown device being evaluated is temporarily replaced by a second horn connected

to a precision receiver by a length of 50 ohm coaxial cable and this receiver must also

present a good 50 ohm termination to the cable. Certain spectrum analyzers and precision

receivers present a better match than others and this should be a major consideration in

selecting one. Now, the field at the receiving site can be calculated from the known waveform

generator spectrum and the transmitting function of the horn. But it can also be measured

by the standard receiving system and the values determined by each method should correlate.

Differences greater than a few dB usually indicate spurious reflections on the range.

To determine the field intensity at any distance along the antenna boresight when the

generator produces a sinusoidal voltage, simply multiply the generator output voltage into

a 50 ohm load by the transmit transfer function and divide by D, the ratio of the distance

in meters r to a reference distance of 1 meter, or

E = VT^/D.

h

Note that D is dimensionless and T, has dimensions of volts per meter/volt. This is valid

only for those values of D that ensure far-field conditions. When this is expressed in dB, as in appendix B, simply add the generator voltage in dB above 1 volt and subtract the distance ratio in dB from the transfer function in dB giving the field strength in dB above

1 volt/meter,

^dBV/m "

'^dBV

"^ \dB/m ~ °dB '

The same equation is used when the input is an impulse, and its spectrum (S, ) is given in

dB

dB above 1 volt-picosecond as is the case for Andrews' impulse generator. We define spectrum

of the field (E',„) as

dB 45
^dBVps/m ^dBVps

"*" \dB/in °dB ' (24)

where £"!„„ , has dimensions of dB above 1 volt-picosecond per meter, and S ,„„ is the dBVps/m dBVps

spectral intensity of the impulse generator in dB above 1 volt-picosecond.

If the spectrum of the input waveform is given in microvolts per megahertz (yV/MHz),

the numbers will be exactly the same as above since

1 pV/MHz = 1 V-ps ,

and the field spectrum will be given in dB above 1 mV/MHz per meter. For example, figure 33 shows that the Andrews' impulse generator has a spectrum of 66.4 dB above 1 V-ps in the

vicinity of 1 GHz. Using horn 1 to transmit. Appendix B shows its response at 1 GHz to be

-8.1 dB at a distance of 1 m. At 40 cm separation, the signal would increase by a factor of 2.5 or 8.0 dB. The field spectrum from horn 1 at 40 cm would then be

E' = 66.4 dB + (-8.1 dB) - (-8.0 dB) = 66.3 dB ^^=^ . aB m

The receiving characteristics can be used to check this field spectrum value. Suppose

horn 3 was located 40 cm from horn 1 which was transmitting the signal described above. The

voltage that horn 3 should supply to a 50 ohm receiver is simply the electric field times

the receive transfer function

V. =ER^ R

or in dB

\dB ^dB "^ ^dB •

Since the receiving function is measured with respect to the field at the receiving

antenna aperture, there is no distance adjustment. For a broadband signal, the equation

gives the spectrum of the received voltage waveforms (S_ ,„) as RdB

S„,^ = E' +

RdB ^dB "^ ^dB • (25)

Appendix B shows the receive function of horn 3 at 1 GHz to be -35.0 dB with respect

to 1 V/— and so the received spectrum should be m

47
This can be measured using a spectrum analyzer as a receiver. The results of this

measurement yielded 28 dB on a 1 MHz bandwidth when the proper corrections were applied.

Sd^h = S ,_ - BW,„ + 107 dB + 3 dB RdB mdB dB

= - 79 dB - 3 dB + 107 dB + 3 dB

= + 28 dB yV/MHz

where

S ,„ is the measured spectrum in dBm mdB

BW is the ratio of actual bandwidth to 1 MHz

uB

107 dB is used to convert dBm to dByV

3 dB is added to convert RMS to peak volts.

The same spectrum measured on the APMS resulted in a figure of 30.5 dByV/MHz and on a commercial EMI receiver using only handbook calibrations, 28 dBpV/MHz.

It is interesting though that the signal is only 4 dB above the noise in the EMI receiver having a bandwidth of 500 kHz; 10 dB above the noise for the spectrum analyzer

with 1 MHz bandwidth, or 20 dB for 3 MHz. The APMS was still making valid measurements

40 dB below this level and extended to 50 dB when the integration time was increased to

1 minute (the repetition rate for the impulse generator was 50 kHz for all measurements).

This minimum equivalent input to the APMS was -20 dBuV/MHz or 0.1 yV/MHz. The increased

sensitivity is due to the signal processing that the APMS can do when it has the prior

knowledge of when the signal will occur.

The received signal strength diminishes rapidly as the antennas are separated, and the

signal will be lost in the noise at a 1 meter spacing if the EMI receiver is used. The

spectrum analyzer, when used on the 3 MHz bandwidth will give the same results with the

antenna spacing increased to 3 meters. The superior bandwidth of the APMS allowed excellent

measurements to be taken at 3 meter spacing and should still operate well at over 100

meters if reflections could be controlled.

In view of the results obtained in the near-field measurement, it is recommended that

the horn antenna pair be used with spacings between one meter and three meters. For a horn

transmitter and some other receiving antenna, the spacing may have to be increased because

of the possible near-field effects due to the receiver antenna.

8. CONCLUSIONS

A method for measuring antenna transfer functions in the time domain was developed, and

the technology is now available to measure the transmitting and receiving functions of most

antennas. Given this capability, the receiving characteristic of a variety of antenna types was measured, and the TEM horn was selected as the most promising candidate for a transfer

standard antenna.

48
TEM horn characteristics were measured on a ground plane and in a free-space field

environment. Preliminary measurements indicate that the receiving characteristics are constant within ± 10 dB from below 100 MHz to above 6 GHz but have only been measured

accurately (± 3 dB) between 600 Mhz and 5.5 GHz. Transmission characteristics were shown to include a 6 dB per octave frequency dependence in accordance with accepted reciprocity relations. This limits the useful transmitting range using impulse excitation to 600 MHz - 5 GHz with a variation from flatness of

less than ± 10 dB. However, a flat transmitted field spectrum was obtained when the

driving signal was a square wave. Using this signal source, the generated impulsive fields

have a reasonably flat spectrum even below 100 MHz. The limits of error have been estimated to be less than ± 3 dB in the frequency range

600 MHz to 5 GHz as long as the antenna spacing is approximately that of the measurements

reported herein, 2,5 meters. The effects of changes in spacing included those due to reflections and near field and did not exceed ± 3 dB over the range 40 cm to 320 cm. Care

has to be taken to minimize the effect of reflections from the ground surface and large

reflecting surfaces close to the antennas. Using baluns with these antennas added an additional uncertainty of ± 3 dB below 1 GHz and ± 2 dB above 1 GHz. Feasibility tests have been made with balanced (push-pull output) generators and

resistively loaded horns which indicate that much flatter responses and significantly

smaller errors can be obtained over a wider frequency range (perhaps ± 3 dB over 100 MHz to

6 GHz).

The preliminary experiments just mentioned indicate that we have good understanding of

those factors which must be changed in the TEM horn to extend its frequency both downward

to 20 MHz and upwards to at least 8 GHz. The flatness can be significantly improved over

this range and, therefore, the sources of errors can be better controlled.

Time did not permit the completion of the Singularity Expansion Method (SEM) and

homomorphic analysis mentioned previously, which would be needed to extend the frequency

range of the ground plane downward, but even without SEM and with some modifications to the

ground plane, it can be used down to 50 MHz. SEM type analysis is expected to extend that

downward to 20 MHz and perhaps lower.

The lower frequency work will probably require removing the measurements to the outdoor

antenna range which will reduce reflections and allow greater antenna spacing. This had to be postponed for later work, lacking the necessary modifications to the APMS to protect it

from the outdoor environment. This is still an important improvement to consider for future

work.

49
3
APPENDIX A

In this appendix the functional relation between the TEM horn transmitting function,

T , and receiving function R^ will be derived. As has been noted previously [4,5], these

functions are not identical functions of time (frequency) . What is the same at a given

frequency and to which the usual reciprocity relations refer is directivity or antenna

patterns.

Martins and Van Meter [5] have given a qualitative relation between the electric field

generated by a horn and its input voltage, the former being the derivative of the latter,

but impedance details are suppressed in their analysis which prohibits one from making a quantitative evaluation.

To obtain the desired result, we will start with a specific example, a uniform sphere

and calculate the on axis transmitting function, T(a),r), and receiving function, RCcd) . This will then be compared to results we will obtain from equations applicable to an arbitrary

antenna. Readers who are interested only in the general result, can proceed directly to the

analysis that begins after eq. (A27).

The magnetic field radiated by a gap fed perfectly conducting sphere [6] , as shown in

figure 43 , is

wea^^V 00 (2n+l)p-'-(o)p-'"(cose) H^^,^ (kr) „ / s o n n n-Ha /ai\ %(-'^) = -^ -^T \ ^^^i^) v„(2),^, „(2)., ^^^

r n=l kaH , (ka) - nH ,, (ka) n-% n-Hs

where

e = The medium permittivity

r = the distance from the center of the sphere to the point of observation

a = the radius of the sphere

k = 2./^

V = the voltage at the gap

P (cose) = the associated Legendre function of the first kind

(2) H (ka) = the Hankel function of the second kind

n which kind is a result of an assumed, but suppressed,

time dependence of the field of e . This results

in a traveling wave in the +r direction.

The transverse component of the electric field, E. (r,aj,e), in the far field can be

9 obtained from eq. (Al) by the usual relation,

(A2)

Eg(r,w,e) = nHg(r,a),9).

The radial function of eq. (Al) in the far field kr>>l, becomes [14]

\^h^^^^ ^^2^1 .n-1 -ikr (A3) U— = V — w i e

z r

TT kr

51
I ^/ ; ^9

^y

\\\

n/2

N

Figure 43. Magnetic field radiated by a conducting sphere.

52
Also, on axis P^(cos 0) becomes P-^(o) which, together with eq. (A3) in eq. (Al) and with

the result In eq. (A2) , one has

nV , ,„ rr- -Ikr

TT, "^ o l/lJzke

E(co,r,-T-) = —;r- wea \- 2' 2 IT kr

I

i"(2n+l) ^^n(°>^'

(A4)

n^l "^^"^^ [kaH^^^ka) - nH^l/(ka)] n-% n+Js

The quantity, P^o) , Is given on page 554 of [2] as n

pl(o) = ^^

r—

^

for n odd, and

^^^^ 2^-^(^)'.]2

= for n even.

Now for the Inverse problem, consider figure 44 which represents a solid metal sphere

embedded in a plane wave field with the polarization and propagation directions indicated

by the E and k vectors, respectively. We will solve for the resultant current across the

equator at (j) = ± y. This will be the desired load current, and it will be assumed to be

the same as the current, I, that would flow in a real, thin, circular load resistor placed

at this equator as long as the conductance, G , across this resistor (in the (j) direction)

is much greater than the spherical antenna source admittance (a typical value is .004 ohms

for ka = 0.5 [6]).

The voltage, V , developed across this conductance then will be

Li

V = ^ . (A6)

If we let a and a represent unit vectors in the r and 6 directions, respectively,

then the current density, J(G), at the equator is

J(e) = a X H(a,6,<())

and the total current, I , is obtained by integrating along the equator or

(A7)

TT _ O _

I^ = a / J(e)-a,de| + a / J(e)-a d(-e)|

L <f . . T^ <^ . '^

(A8) = a / [H(a,e,y) + H(a,e,- y)]de,

The required expression for H(a,9,())) can be obtained from Stratton [15]. For the

problem of interest, the vector function m-. will have only a 6 component since

cos(± tt) = 0. The vector function n , will not exist since cos(± -;r) = eliminates the °^" 3pl(cos 6)

2

53
V

;2r=+n/2

1 —^y

Figure 44. Conducting spherical receiver in a plane wave field.

54
component, integration of the function, r— , over all 9 eliminates the 9 component,

ad and continuity of the magnetic field at a perfectly conducting boundary requires that the r component be zero.

The explicit result for the sum of the incident, H. , and reflected, H , radial components r ir rr

of the magnetic field is [15]

H. + H = f (ca,9,(t))[n^^^ + a^ n^^-* ] • a

= f'(a.,9,^)[j^(p^) - \° "

] (A9) n o = 0,

where p = ka, f(a),9,({)) and f ' (o), G,(t)) are arbitrary functions independent of r, and j (p)

and h (p) are spherical Bessel and Hankel functions related to the more conventional

functions by

„(p) =*/-^- J ^ (P) (AlO)

2kp

and

We are thus left with a single 9 component for m which yields for the magnetic field

kE oo .n,_ ^, . pl(cos 9) H. = -^ I M%7V- "

, sin 9 j (p )a, (A12)

1 yo) ^T n(n+l) sin 9 -^n o 6 n=l

where the assumed time dependence, e , has been suppressed. Now for the total field

H. +H =^ I ^^^^^ (m(|> + b^ m(,3>) 1 r yw ^, n(n+l) eln n eln n=l

(A13)

kE °° .n,„ j^.

o

Y 1 (2n+l) yu ^T n(n+l) n=i

pl(cos 9) f [Pj„(p)]'

3„(p) - sin 9 ' " [ph^^^(p)]' , n

^'^'' P=Po

where the primes denote differentiation with respect to p . In particular,

[Pjn(p)]' = J^(P) + P f^ [j^(p)] (A14)

and

[ph^^^p)]' = h^^p) + p^ [h (p)]. (A15) n n dp n

55
Therefore, the quantity In the braces becomes

jjp) h^i)(p) + pj^(p)^ [hf )(p)] - jjp)h;;i>(p) - Ph^^^p)f^ lijp)]

= p

^r<p>-»i(H^"(p)i

Jn'-'^t'-n"'^)' -^u'('>h [J„<°)1

P=PO (A16)

"'"'P'-^il-r'^)! P=PO

Now the quantity in the numerator is simply the Wronskian of j (p) and y (p) times i or if—T-] , To demonstrate this we will substitute

h^^^p) = j^(p) + iy^(p)

in the expression of interest, eq. (A16) , which gives [14]

^n(P)fef^n(p)]+ii[yn(P)]' 'dp "n(P)MJn(P> -^ ^^n^P)

= ^K^^^ i f^n^^^^J - ^n(P) i [Jn(^>] (A17)

The denominator of eq. (A16) can be expressed in an alternate form by using the recurrence relation [14]

^hil)(p)+f-[h(l)(p)]=hil)(p)

p n dp n n-1 (A18)

which together with eq. (A17) in the expression of interest, eq. (A16) , gives

i/p?

p h (p ) - nh (p ) o n-1 o n o

(A19)

for the term in curly brackets, which inserted in eq. (A13) and with that result in

eq. (A8) one obtains for the current

IT pl (cos e)

9 r -^ Hfi

I (A20) = ^ Ly]?^ y i"(2n+l)

n=l n(n+l) [kaH^\^(ka) -nH^^Cka)] n~^ TX'T^

Note: If a wave of time dependence, e , had been chosen (as was done in deriving eq.

(2) (A4)) the Hankel function would have been of the second kind, H ,, (ka) , as can be seen from

"^"^

( 1

10.1.17, page 439 of [14] which yields e times a polynomial in 1/z for h (z) and requires

56
e to generate a traveling wave. Note also that the sums in eqs. (A4) and (A20) are proportional since

TT P^ (cos 6) ^. «, >, 2 / " . , de = 2iT[pl(G)]2 (^21)

sin G n

o

1 which can be seen to be true by writing out the polynomial which represents P (cos 6) and

performing the operation indicated above term by term. Now for a sphere that is small compared to a wavelength, the antenna input admittance in

the transmitting case can be much less than the characteristic admittance of the guide which

feeds the antenna. In this case the net voltage, V , into the load is ° o

V = 2V. (A22a)

o 1

TT Therefore, the transmitting function of the sphere, T (u),r,y), which is given by eq. (A4)

divided by V., can be expressed as a function of the receiving function of the sphere, R (w)

,

which is given by eq. (A20) divided by G E . In other words,

J-i o

I —ikr ^ T(i.,r,h = mey/^\— I ... (A22b)

n=l

and

R (,) =i2^V^ I ... (A23) L ' n=i

Combining eqs. (A22) and (A23) yields

• n 2 ~ik^

^s^'^'^'f) = 27^]^ ^s^'^)- (^24)

This equation can be simplified using the identity,

(u ye _ k_

27rk 2Tr

^ 1

(A25)

Therefore,

T^(a),r,|) = inG^ j^ e

^^'^ R^(w) (A26)

is the required relation between the transmitting and receiving functions.

The broadband implication of

portional to wavelength or, since

The broadband implication of eq. (A26) is that the ratio of T to R is inversely pros s

2Tri _ iw (A27)

A ~ c '

57
the impulse response in the time domain corresponding to T is the time derivative of the

impulse response corresponding to R . The above result has been derived for the specific case of a spherical antenna which

could be solved in detail and for which we could see the effect of all the parameters. We

will now show this result; namely that the transmitting transfer function is the time

derivative of the receiving transfer function, is more general, and in fact can be obtained

from general reciprocity relations

.

Our equation for T, (u) can be written in a more general on axis form from eq. (1.61)

of Kerns [16] as

\^^'^) (A28) ^^'^'^^ -

a (o)) o ikr

= -ik S,^(o)

^ -10'^ r

= k = (jjVpe and R = on axis.

The reception function into a matched load is found from eq. (1.6-9) of [16] to be

b (u)

(A29)

where A on axis is given by

E(a))

A = 2TrE(a))e . (A30)

Therefore

""ilcir R^Cto) = SQ-|^(o)2Tre on axis. ^^^^

This can be recast in terms of S^„(o) by the receprocity relation, eq. (1.6-20a),

which is

since y = k. With eq. (A32) in eq. (A31) we obtain, at the reference plane,

R^(w) = S^q(o) -^ 277. (A33)

58
We can now obtain the generalized result for the ratio of T^(w) to R^(a)) by dividing

eq. (A28) by eq . (A33) which gives

Tj^Coi.r) -ikS^Q(o)e'-^''/r

P o„ (A34)

S^o(o)

^ 2.

o

o 1 ikr

' II

Xr

^

This result is identical with eq. (A26) for the sphere when one identifies the matched

waveguide characteristic^ admittance, t] in Kern's notation, with the load conductance, G , O i-i for the sphere and 1// — with the impedance, n, of free space.

This then yields the derivative relation between on axis, far-field radiated fields and

plane wave receiving functions, but it is no longer restricted to a specific type of antenna.

The waveguide used for all work reported was coaxial transmission line of Z impedance. When

this is substituted into eq. (A34), the final form of the equation becomes,

T. (aj,r) . .,

I

59

Hi
APPENDIX B

Responses of Army Horn Antennas (Serial Nos . 7-76-1 and 7-76-3)

in Both Receive and Transmit Modes With Appropriate Baluns

Freq. MHz

600

700

800

900

1000

1100

1200

1300

1400

1500

1600

1700

1800

1900

2000

2100

2200

2300

2400

2500

2600

2700

2800

2900

3000

Receive

Horn 1 dB

-34.2

-35.2

-35.5

-35.2

-35.4

-34.9

-35.3

-35.3

-35.7

-35.7

-36.5

-36.9

-36,6

-37.1

-37.4

-36.6

-36.8

-37.3

-37.0

-37.1

-36.5

-36.6

-37.0

-36.7

-36.9

Receive

Horn 3 dB

-34.8

-34.6

-35.5

-35.6

-35.0

-34.9

-35.4

-35.6

-35.6

-35.6

-37.6

-38.2

-37.5

-37.8

-37.7

-37.0

-37.3

-37.4

-37.3

-37.4

-37.3

-37.0

-37.5

-37.8

-36.9

Transmit Transmit

Horn 1 Horn 3 dB dB

-10.4 -11.7

-10.5 - 9.8

-10.0 - 9.9

- 8.6 - 9.0

- 8.1 - 7.4

- 6.2 - 6.5

- 5.8 - 5.7

- 5.3 - 5.6

- 4.7 - 4.7

- 3.9 - 4.0

- 4.2 - 5.8

- 4.4 - 5.8

- 3.7 - 4.3

- 3.5 - 4.2

- 3.6 - 3.6

- 2.3 - 2.6

- 1.8 - 2.6

- 2.1 - 2.0

- 1.1 - 1.3

- 1.1 - 1.2

- 0.4 - 1.2 + 0.1 - 0.1 + 0.3 + 0.1 + 0.8 - 0.4 + 0.6 + 0.8

60
Responses of Horns 1 and 3 Continued

Receive Receive

Freq. Horn 1 Horn 3 MHz dB dB

3100 -36.9 -36.4

3200 -36.4 -37.1

3300 -36.5 -36.8

3400 -36.1 -35.6

3500 -35.9 -36.3

3600 -36.4 -36.7

3700 -36.5 -36.0

3800 -36.4 -36.6

3900 -36.4 -37.7

4000 -36.7 -37.4

4100 -37.6 -37.3

4200 -37.7 -38.3

4300 -38.1 -38.4

4400 -38.0 -37.3

4500 -37.7 -38.3

4600 -38.0 -39.0

4700 -38.4 -37.9

4800 -38.4 -38.4

4900 -39.0 -40.1

5000 -39.6 -40.0

5100 -40.5 -40.1

5200 -40.8 -42.1

5300 -41.3 -41.8

5400 -42.7 -41.3

61

Transmit

Horn 1 dB

Transmit

Horn 3 dB + 1.1 + 1.7 + 2.1 + 1.4 + 2.0 + 2.0 + 2.6 + 3.1 + 3.2 + 2.9 + 3.0 + 2.7 + 3.0 + 3.6 + 3.4 + 3.4 + 4.0 + 2.4 + 3.6 + 2.9 + 2.9 + 3.5 + 3.3 + 2.8 + 2.4 + 2.3 + 2.6 + 3.4 + 4.2 + 3.3 + 3.7 + 2.8 + 3.0 + 3.8 + 3.7 + 3.8 + 3.7 + 2.6 + 3.0 + 2.6 + 2.1 + 2.6 + 2.3 + 0.9 + 1.6 + 1.3 + 0.5 + 2.0

Ifii
REFERENCES

[I] Baum, C. E. , Emerging technology for transient and broadband analysis and

synthesis of antennas and scatterers, Proc. of IEEE, 64, No. 11, 1598-1616

(Nov. 1976).

[2] Susman, L. , and Lamensdorf, D. , Picosecond pulse antenna techniques, Rome

Air Development Command Technical Report RADC-TR-71-64 (1971).

[3] Fitzgerald, D. J., and Martins, V. C. , Impulse response measurement probes

and instrumentation, Rome Air Development Command Technical Report,

RADC-TR-74-288 (1974).

[4] Mayo, B. R. , Howells, P. W. , and Adams, W. B., Generalized linear radar

analysis. Microwave Journal, 4_, 79 (1961).

[5] Martins, V. C. , and Van Meter, J. L., Picosecond pulse reflector antenna

investigation, Rome Air Development Command Technical Report,

RADC-TR-73-215 (1973).

[6] Ramo, S. , Whinnery, J. R. , and VanDuzer, T. , Fields and waves in

communication electronics, (J. Wiley and Sons Inc., New York , N.Y.,

1965).

[7] Cans, W. L. , and Andrews, J. R. , Time domain automatic network analyzer

for measurement of rf and microwave components. National Bureau

of Standards Technical Note 672 (Sept. 1975).

[8] Cans, W. L. , Present capabilities of the NBS automatic pulse

measurement system, IEEE Trans, on Instr. and Meas., IM-25 , No. 4,

384-388 (Dec. 1976).

[9] Andrews, J. R. and Cans, W. L. , Time domain automatic network analyzer,

L'onde electrique, 55^, No. 10, 569-574 (Oct. 1975).

[10] Riad, S. M. , The theory and application of the homomorphic transformation

to time domain spectroscopy and scattering problems, Ph.D. Dissertation,

Univ. of Toledo (Aug. 1976).

[II] Kanda, M. , A Relatively Short Cylindrical Broadband Antenna With Tapered

Resistive Loading for Picosecond Pulse Measurements, IEEE Trans. Ant. & Prop.,

Vol. AP-26, No. 3, 439-447 (May 1978).

[12] Andrews, J. R. , Proceedings of the 1975 IEEE Symposium on EMC (Oct. 1975).

[13] Andrews, J. R. , and Arthur, M. G. Spectrum amplitude, definition, generation

and measurement, NBS Technical Note 699- (Oct. 1977).

[14] Abramowitz, M. , and Stegun, I. A., Handbook of mathematical functions

(Dover Publications Inc., New York, N.Y., 1965).

[15] Stratton, J. A., Electromagnetic theory (McGraw-Hill, New York, N.Y. 1941).

[16] Kerns, D. M. , Plane-wave scattering theory of antennas and antennainteractions: formulation and applications, J. Res. Nat. Bur. Stand.

(U.S.), 80B (Math. Sci.), No. 1, (Jan-Mar. 1976).

62
T14A (REV. 7-73)

U.S. DEPT. OF COMM. BIBLIOGRAPHIC DATA

SHEET

1. PUBLICATION OR REPORT NO.

NBS TN-1008

2. Gov't Accession

No.

3. Recipient's Accession No.

4. TITLE AND SUBTITLE

Antennas and the Associated Time Domain Range for the

Measurement of Impulsive Fields

5. Publication Date November 1978

6. Performing Organization Code

724.04

7. AUTHOR(S)

-R. A. Lawton and Arthur R. Ondrejka

8. Performing Organ. Report No.

9. PERFORMING ORGANIZATION NAME AND ADDRESS

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

WASHINGTON, D.C. 20234

10. Project/Task/Work Unit No. 7244153

11. Contract/Grant No.

12. Sponsoring Organization Name and Complete Address (Street, City, State, ZIP)

Same as Item 9.

13. Type of Report & Period Covered

14. Sponsoring Agency Code

15. SUPPLEMENTARY NOTES

16. ABSTRACT (A 200-word or /ess factual summary of most si^ificant information. If document includes a significant

bibliography or literature survey, mention it here.)

This report describes the construction and evaluation of a TEM horn antenna

designed at NBS to be used as a transfer standard to generate and measure impulsive

electromagnetic fields. Our purpose in the evaluation was to analyze the different

electrical field generation and measurement techniques thoroughly enough to

determine the major sources of error and establish a standard of impulsive field

strength having a well established statement of accuracy.

The evaluation of this horn was done in two independent ways; by placing the

horn in the field of a conical transmission line and by a three antenna intercomparison. The two methods were found to agree within ± 3 dB over the range of

0. 6 to 5 GHz. Part of this disagreement is due to the assumption of far field

conditions, and an experimental technique is described which determines the

frequency range over which this assumption is valid.

17. KEY WORDS (six to twelve entries; alphabetical order; capitalize only the first letter of the first key word unless a proper name; separated by semicolons)

Antenna; conical transmission line; impulsive fields; impulse response; standards;

TEM horn antenna.

18. AVAILABILITY g^ Unlimited

I ! For Official Distribution. Do Not Release to NTIS

ly I Order From Sup. of Doc, U.S. Government Printing Office Washington, D.C. 20402. KKX^OLXHaCXXH

I 1 Order From National Technical Information Service (NTIS)

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19. SECURITY CLASS

(THIS REPORT)

UNCLASSIFIED

20. SECURITY CLASS

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21. NO. OF PAGES

68

22. Price

$2.40

tt U.S. GOVERNMENT PRINTING OFFICE: « 978-0-677-07 2/ 1 209 USCOMM-DC 29042- P74

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