NATL INST. OF STAND & TECH 5urement Services:

^ f REFERENCE FUBifCAiioNs

Calibration Service of Optoelectronic nist

Frequency Response at 1319 nm for pSbuciti^^

Combined Photodiode/RF Power 250-51

Sensor Transfer Standards

Paul D. Hale and C. M. Wang

IC U.S. Department of Commerce

Technology Administration

00 National Institute of Standards and Technology

157

10.250-51

I 999
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NIST Special Publication 250-51

NIST MEASUREMENT SERVICES:

Calibration Service of Optoelectronic

Frequency Response at 1319 nm for

Combined Photodiode/RF Power

Sensor Transfer Standards

Paul D. Hale and C. M. Wang

Optoelectronics Division

Electronics and Electrical Engineering Laboratory

National Institute of Standards and Technology

325 Broadway

Boulder, CO 80303-3328

December 1999

U.S. Department of Commerce

William M. Daley, Secretary

Technology Administration

Dr. Cheryl L. Shavers, Under Secretary of Commerce for Technology

National Institute of Standards and Technology

Raymond G. Kammer, Director
National Institute of Standards and Technology Special Publication 250-51

Natl. Inst. Stand. Technol. Spec. Publ. 250-51, 51 pages (Dec. 1999)

CODEN: NSPUE2

U.S. GOVERNMENT PRINTING OFFICE

WASHINGTON: 1999

For sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, DC 20402-9325
Contents

1. Introduction , 1

2. Principle of heterodyne frequency response measurements 2

2.1 Photodiode with access to bias monitor 3

2.2 Combined photodiode/rf power sensor transfer standard 5

3. Nd:YAG heterodyne measurement system 6

3.1 Basic setup and swept operation 6

3.2 Phase-locked operation 11

3.3 Power matching 12

4. Uncertainty in combined photodiode/power sensor normalized response 13

4.1 Type A uncertainties 13

Repeatability 13

Ofl&et correction 17

Environmental conditions 19

4.2 Type B uncertainties 24

Meter range scaling 24

Meter offset 24

Optical power drift (power matching) 24

Bias current measurement 24

Power sensor noise 24

Reference power uncertainty 25

Gunn oscillator frequency drift 25

Hysteresis 25

Relative intensity noise 25

5. Quality Control 26

5.1 Calibration notebook , 26

5.2 Checks on each calibration 26

5.3 Periodic checks 26

Power stability 26

Gunn oscillator frequency 26

Check standards 26

5.4 Future system changes 27

5.5 Transfer standards which are suitable for calibration 27

6. References 27

iii
Appendix A: Use oftransfer standard in a ratio system 30

A.l Application ofheterodyne formalism to arbitrary modulation depth 30

A.2 Calibration transfer using ratio measurements 31

A.3 Sources of uncertainty in ratio measurement system 33

Type ofphotoreceiver specimen 33

Spurious harmonics 33

Relative intensity noise 33

Wavelength dependence 33

Reference 33

Appendix B: Heterodyne system with unequal powers 35

Appendix C: The meaning ofnormalized response and 0 dB 36

Appendix D: Photoreceiver with no access to bias monitor 37

D.l Expression ofabsolute responsivity in terms of measured quantities 37

D.2 Uncertainties of amplified receiver/power sensor combined response 37

Coupling ratio 38

Receiver linearity 38

Receiver stability 38

Receiver noise 38

Appendix E. Sample calibration certificate 40

Appendix F. Shipping instructions 47

iv
Calibration Service of Optoelectronic Frequency Response at 1319 nm for

Combined Photodiode/rf Power Sensor Transfer Standards

PaulD. Hale and C. M. Wang

National Institute of Standards and Technology

Boulder, Colorado 80303

This document describes the Calibration Service provided by the National Institute

of Standards and Technology (NIST) for optoelectronic frequency response

transfer standards and their use in measuring optical modulation depth. The

service supports the calibration of transfer standards consisting of a pigtailed

photodiode, vsdth access to the bias current, combined with an rf power sensor over the frequency range of 300 kHz to 55 GHz at 1319 nm. Typical

measurement uncertainties are about 0.05 dB, which includes measurement

repeatability, rf power meter scaling accuracy and ofifeet, optical power drift, bias

current measurement, and power sensor noise. The measurement technique, based

on the heterodyne beat note between two single-frequency Nd:YAG lasers

operating at 1319 nm, is reviewed and its implementation is described. A detailed

analysis of sources of error and estimates ofuncertainty are presented. Use of the

transfer standard for measuring an arbitrary optical modulation depth is described

along with some design considerations to achieve the lowest possible transfer

uncertainty. A modification ofthe calibration service that allows calibration of

photoreceivers that do not provide access to the bias current is also described.

This modification, offered as a Special Test, allows the absolute responsivity of the

photoreceiver to be measured, but with degraded uncertainty due to repeatability

of the optical insertion loss of the fiber connectors.

Key words: calibration; frequency response; modulation depth; optical receiver;

photodiode; reference receiver; transfer standard

1. Introduction

Increased commercial availability of optoelectronic devices and test equipment requires the

availability of accurate and inexpensive optoelectronic frequency response calibrations that are

traceable to national standards. Examples of this need are the standard test receivers used for

SDH/SONET (synchronous digital hierarchy/synchronous optical network). Fibre Channel, and

Gigabit Ethernet that are specified [1] as having a fourth-order Bessel-Thompson filter response

with tolerances [2], [3] as low as ±0.3 dB to ±0.5 dB. Instrumentation for analog applications

such as CATV have comparable tolerances. Our customers prefer transfer standard receivers

that are calibrated with uncertainties that are at least a fector of4 smaller. Until recently,

uncertainties this low were not possible[4], [5].

1
In the past, photoreceiver frequency response measurements have had high uncertainties for two

reasons: inaccurate knowledge ofthe optical stimulus and large uncertainties in the microwave

power measurement. With careful attention to system performance, uncertainties around 0. 12

dB to 0.45 dB can be achieved [5]. This accuracy is not adequate to support the ±0.3 dB

SDH/SONET tolerance specifications with adequate confidence.

In a well designed heterodyne measurement system, the variation in optical stimulus can be

reduced to ±0.0023 dB (coverage factor k=\). Typical rf power sensor calibration uncertainties are about a factor of 1 5 or 20 higher (for k= I). Uncertainties due to mismatch can be 0. 1 dB or

higher. Mismatch errors can be corrected, but it is unclear whether the uncertainty in the

corrections is low enough to give adequately improved uncertainty. If the uncertainty due to rf

calibration and impedance mismatch could be eliminated, the overall optoelectronic response

uncertainty could be significantly reduced.

In this work, we describe a method for calibrating a photoreceiver frequency response transfer

standard with the low inherent uncertainty of a Nd:YAG heterodyne system. This is achieved by

combining a photoreceiver with a microwave power sensor and calibrating the response ofthe

combined unit, eliminating rf sensor calibration and mismatch uncertainties. The measured

response is normalized to the dc electrical power generated in the photodiode to give the lowest

possible uncertainty. The Nd:YAG heterodyne measurement system is also described along

with a detailed uncertainty analysis for calibration of a combined photoreceiver/power sensor

transfer standard. The theory for measuring an unknown device against the transfer standard

and measurement ofmodulation depth is also derived.

Calibration of frequency response normalized to the optical power is also described. This type of

normalization gives absolute responsivity, but with a degraded uncertainty, and is especially

suited to calibrations of amplified photoreceivers. It is offered as a Special Test because the

uncertainties due to fiber connector insertion loss can not be fiilly evaluated.

2. Principle of heterodyne frequency response measurements

The advantages and disadvantages ofvarious methods for characterizing photoreceivers have

been discussed by several authors [4], [6], [7]. The main advantages of the heterodyne method

are that the optical modulation is completely calculable from fundamental principles, it gives

results directly in the frequency domain where many industrial specifications are written, and

tuning ranges are available over 65 GHz or more. The heterodyne principle is based on a

property of all optical quantum detectors: the current generated by the detector is proportional to

the time-averaged energy flow (5) integrated over the surface of the detector. Here S is the

Poynting vector and its time average (assuming a monochromatic plane wave) is [8]

is) = -

2M

^

{E-E*)k (1)

0

2
where Eq is the permittivity of free space, |io is the permeability of free space, E is the electric

field vector, E* is its complex conjugate, and A; is a unit vector in the direction of propagation.

The heterodyne beat note at frequency/ is generated by mixing signals from two "singlefrequency" lasers as shown in figure 1. If the laser beams have the same polarization and the

curvature and direction of propagation of the laser beams after passing through the beamsplitter

are the same (the beams are mode-matched) the electric field incident on the detector can be

written as

E(x,y,tME^,e'^' ^ E^,e'^^'^-f^')Fi,x,y) . (2)

Equation (2) can then be integrated over the surface ofthe detector eliminating the spatial

dependence in F/x,y) to give the power incident on the detector. Let the power delivered to the

detector by the rth laser be P^/ ^ ^l^o/l^^ (^(/Mo)''^' where A = j j\F/x,y)\^dxdy integrated over

the area ofthe detector, and Egi is the electric field. Then the total power incident on the

detector is

P^o,ai(t) = iPn, - P02) ^

01 02> yPoiPo2 COS(27t//)

(3)

Here we have assumed that the power from

each laser is constant and that each laser is truly single frequency. The phase relationship

between the lasers has also been neglected.

2.1 Photodiode with access to bias monitor

The photodiode in figure 1 is connected to an

external bias network and dc blocking

capacitor. A dc return path must be provided

for the photodiode, either through a bias T or through a resister or attenuator. The

photocurrent generated by the photodiode due

to the optical power incident on it is given by

Pif) cos(27r/r+(|)(/))

(4)

where R(f) is the responsivity of the detector

(in A/W) at frequency/and (t)(/) accounts for

phase delay in the photodiode, transmission

line, and connectors. For the purposes of this

Beamsplitter

(>

Rl

ntn

Figure 1. Simple photodiode circuit showing

generation of photocurrents.

3
work phase can be neglected. Equation (4) assumes that the detector is reverse-biased and uses

the convention of positive current flowing in the reverse direction ofthe diode. The first term on

the right side is the dc photocurrent, which flows through the bias supply, and the second term is the rf photocurrent, which flows through the rf load (the microwave power sensor) through a dc

blocking capacitor. The mean squared photocurrent generated by the photodiode is

<if)> = (Poi ^ Poif ^'(0) ^ ^Poi Po2 P\f)

2 ^ ^ .2 = <l,> + <l

(5)

If Pqi is nearly equal to Pq2, then 2(Pq,Pq2) is, to first order, equal to 0.5(Pq] + Po2f, as discussed in Appendix B. The normalized fi-equency response U^{f) which is defined as R\fyR\0), can then be found by taking the ratio of the rf electrical power P^ to half the dc

electrical power delivered to a load Ri ,

2^./ _ <^>Pl

<ii>Rr dc 0.5<ii>R ^^L ^"^ 'dc^^L

(6)

2{P^,P^,)R\f)R,

o.5(p^, Po2n\m,

R\f)

R\0)

'

Prfis a function of fi-equency. It includes corrections for the sensor calibration factor and

electrical mismatch, and is the power that would be delivered to an ideal load Rl [9].

Measurements are typically made with transmission lines and loads with characteristic impedance

of 50 Q; however these equations also apply to measurements with 75 Q systems. The

normalized fi-equency response may be quoted in decibels as 201og[Sl(/)]. All NIST calibrations

report the normalized response in 20\og[Bi(f)] format. The electrical bandwidth of the device

is where 201og[3l(/)] falls by 3 dB from the low frequency level [3].

In an ideal measurement < i^^ > is constant, but in any real measurement system it may vary

because of changing optical power coupled to the photodiode. The ratio in eq (6) is insensitive to optical power variations. Using the normalized response also simphfies the measurement

apparatus because only the total photocurrent in the photodiode need be monitored instead ofthe

power coupled to the detector firom each laser.

The normalized fi-equency response is an unbiased method for comparing measurements made

with different systems. A common practice when comparing fi*equency response measurements

is to add an offset to each curve to make the responses equal at some fi*equency or over some

4
range of frequencies. This practice assumes that both systems impart negligible error at that

frequency and can be misleading.

Another method of comparison uses the absolute response. This method suffers from connector

insertion loss reproducibility and optical power uncertainty. The absolute rf responsivity is calculated from the normalized frequency response by multiplying by the square of the dc

responsivity, which is measured separately. Because most fast commercial photodetector

packages are pigtailed and connectorized, the uncertainty of the absolute responsivity includes

the uncertainty ofthe fiber connector insertion loss. Variations in fiber connector insertion loss

may be high and can contribute to significant uncertainties in the absolute responsivity [10], [1 1]. A correction must be made for these variations to attain the desired uncertainty. This problem

will be discussed further in Appendix D and Chapter 4. High speed detectors that are thin may

also have significant wavelength dependence of the absolute responsivity due to interference in

the photodiode.

2.2 Combined photodiode/rf power sensor transfer standard

Measurement of a photodiode' s unknown frequency response using a photodiode (transfer

standard) calibrated using eq (6) includes calibration uncertainty of both the power sensors used

to measure from the photodiode transfer standard at NIST and at the customer's laboratory,

and calibration uncertainty ofthe sensor used to measure the photodiode, giving a total expanded

uncertainty of 0.2 dB or more for the modulation transfer function. Combined with other factory

uncertainties, this may give an unacceptable uncertainty for the intended test system

The transfer uncertainty can be reduced by ceilibrating the frequency response of the photodiode

combined with the rf power sensor and bias T or attenuator used for dc return. This method

totally eliminates uncertainties due to power sensor calibration and impedance mismatch. The

combined frequency response B?(f) measured on the NIST heterodyne measurement system

includes the power sensor calibration factor and impedance mismatch, and is given by

'if) - C

0.5<ii>R,

(7)

In eq (7), P„ is the indication of the power meter after zeroing and calibration against the 50

MHz reference signal (from the power meter) using a calibration factor of 100 %. C is given by

5
^ T^^ ^pd^sensor\^ ' (8)

where k is the sensor calibration factor and Tp^ and r^g„^^^ are complex reflection coefficients.

However, C does not need to be known and the power meter reading does not need to be

corrected with a frequency dependent calibration factor. The form of C is stated here only for

completeness.

The photodiode, attenuator or bias T, and rf power sensor can be left free to be disconnected at the plane of the power sensor connector, or they can be permanently joined. Each of these

methods has potential trade-offs. If the power sensor is free to be disconnected, the internal gain

ofthe power meter can be scaled according to the 50 MHz reference source on the power meter.

This may be an advantage if the transfer standard will be used to measure modulation depth (see

Appendix A). However this introduces uncertainty due to the disconnect/reconnect procedure

and can be operator dependent. This uncertainty is a fimction of frequency.

If the power sensor is kept attached to the photodiode, the internal gain cannot be scaled to the

50 MHz reference source, and long-term drifts in the gain can lead to offsets (on a logarithmic

scale) in the response. These offsets can be assumed to be independent of frequency since the

measurement is made over a short time relative to the drift. Then the offsets can be corrected

using a least-median deviation, as described in Chapter 4. This type oftransfer standard

calibration would be preferred for measurements that are more concerned with the frequency

response at one frequency relative to another frequency than with the "absolute" normalized

frequency response. For example, this type of transfer standard could not be used to measure the

actual fractional modulation depth of a source, only changes in the modulation depth with

frequency (see Appendix A). The choice between these methods is left to the customer.

3. Nd:YAG heterodyne measurement system

3.1 Basic setup and swept operation

The Nd:YAG heterodyne system uses two commercially available single-mode monoUthic-ring

Nd:YAG lasers operating at 1.319 [im. The system of lasers, mirrors, attenuators, acousto-optic

modulator (AOM), and beamsplitters is shown in figure 2. The laser YAG3 is a new high-power

model that replaced the laser previously called YAGl , The beam from YAG2 travels through the

AOM and is combined with the beam from YAG3 in beamsplitter BS 1 . The combined beams

then go through polarizing isolators. These reduce the effects of optical feedback on the lasers

and ensure that the polarization of each laser is the same before entering standard single-mode

fiber through variable focus collimators. The collimators have return loss greater than 30 dB to

reduce multiple path interference in the fibers. The AOM uses a traveling acoustic wave to

diffi-act a small fraction of the beam from YAG2. This beam is Doppler-shifted down in

frequency by 70 MHz and is combined with a small portion ofthe beam from YAG3 in

6
Intensity control

loop

Intensity control

loop

—p. to fast detecter,

J

counter, and PLL

70 MHz Osc

Detector to PLL

(Figure 4)

To specimen, spectrum

aniayzer, and optional

power meter

Figure 2. Schematic of heterodyne measurement system. Labeled components are beamsplitter

(BS), variable beamsplitter or variable attenuator (VBS), mirror (M), acousto-optic

frequency shifter (AOM), neutral density filter (ND), half-wave plate (HWP), isolator

(I), and fiber-collimator (FC).

beamsplitter BS2. The resultiag beat note, which differs in frequency by 70 MHz from the main

signal at BSl, may be used for phase locking the lasers and controlling their frequency difference

with a frequency-synthesized signal generator. This option will be discussed in section 3.2.

The frequency ofeach laser can be tuned thermally to give beat frequencies from nearly dc to

greater than 55 GHz; the beats have a short-term bandwidth of less than 3 kHz. The beat

frequency is measured with a microwave counter (see figure 3). As the frequency is swept, data

are acquired automatically. The data spacing is usually selected to be one-tenth to one-fifth of the

customer's requested data spacing. The frequency resolution of the system is limited by the

sweep rate, the frequency jitter, and the time constants of the data acquisition equipment. At

7
I Optical Input

From Lasers

I

3 dB Fiber

Coupler

1 GHz

Amplified

Receiver 5 GHz Amplifier

2.92(f) - 2.4(m)

Adapter ,

Input 2

Input 1

40 GHz

Detector

^nrHr 2.4(m)- 3.5(f)

Amplifier Adapter

Counter

Figure 3. Frequency counter subassembly for operation below 40 GHz. Although input 1 is specified between 500 MHz and 40 GHz, it does operate, with some cut-outs, below

500 MHz. Input 2 operates between 10 Hz and 80 MHz in high Z (1 MQ) mode and

between 10 MHz and 525 MHz in 50 Q mode. The 3 dB coupler can be omitted and

the optical input can go directly into the high bandwidth detector for high bandwidth

scans where the detector sensitivity is degraded.

present, the highest resolution achievable in swept mode is about 300 kHz, although the

resolution may be considerably worse when the frequency is swept quickly (for example when a much larger data spacing is desired). Higher resolution can be achieved using the phase-locked

loop described later in this chapter. Since the frequency range ofthe rf counter is limited to

about 40 GHz, an external mixer is required to down-convert frequencies above 40 GHz to a

range where the counter operates. The circuit for down-conversion of frequencies in the 40 to

55 GHz range is shown in figure 4. The beat signal is generated in a 40 GHz bandwidth detector

and amplified in a 50 GHz bandwidth amplifier. This signal then enters the rf port ofthe mixer.

The LO (local oscillator) is provided by a Gunn oscillator operating at about 62.15 GHz. The

down-converted signal from the IF port is then amplified with a 26 GHz bandwidth amplifier and

fed into channel 1 of the rf counter.

8
50 GHz

Amplifier

1.85(m) to WR-1 9 Adapter

62.15 GHz

Gunn Oscillator

2 92(p - 2.4(m)

Adapter

40 GHz

Detector ^ Optical Input

from Lasers

40-60 GHz Mixer

26 GHz

Amplifier

33 GHz Bandwidth

Coaxial Cable

40 GHz RF Counter

Input 1

Figure 4. Subassembly for measuring frequency between 40 GHz and 58 GHz. System actually

operates down to about 36 GHz.

Three generic types of open loop sweeps can be performed. The first is a scan of up to 40 GHz

in 2 parts through a mode-hop. The temperature of both lasers is initially set so that the

difference frequency is about 40 GHz. This positions each laser at the extreme end ofthe

temperature range where single-mode operation is possible. YAG2 is then scanned toward the

opposite end of its stable region ending with a difference frequency aroimd 18 GHz. YAG3 is then shifted by about 20 GHz, and YAG2 is tuned back to where it started. At this point VBS2

must be manually adjusted to compensate for the changed power from YAG3 that is coupled into

the fiber going to the photodiode specimen. We believe that the direction ofthe beam from

YAG3 changes slightly when the laser's temperature changes, causing the power that is coupled

into the fiber to change. If YAG2 is ever replaced with a similar new laser, this problem must be

corrected. YAG2 is then scanned as before with a starting frequency near 20 GHz and sweeping

through dc. When diode power sensors are used, the sweep is usually done too quickly to

reliably change input ranges on the counter to measure frequencies below 500 MHz, causing

some drop-outs in that region. The finest sample spacing achievable without using an external

9
Optical signal

from BS2

1 GHz receiver BiasT

Monitor

RF

Synthesized signal

generator

8 dBm

PLL box detailed in Figure 6

LO

RF

Fast Out

on/off

Power

supply

i 7

£ Manual gain

adjust

YAG3 piezo input

20 second

time constant

YAG2

1 .5X;>—^ thermal

tuning

1X

^YAG3 -"^^ thermal

tuning

Figure 5. Schematic of circuit for frequency control, including phase-locked operation. Power

supply for PLL box is turned off for swept (open loop) operation.

voltage divider is about 5.8 MHz using the 1.5x external amplifier, giving the finest calibration

data spacing of about 45 MHz.

The second type of scan is designed for finer resolution scans and can automatically switch

between counter inputs to select the correct operating frequency range. YAG3 is left at one

position and YAG2 is swept to achieve sample spacing as low as 30 kHz. The finest frequency

resolution attainable in an open loop or swept mode can be achieved by attenuating the input to

the external amplifiers for YAG2 using the 1.87kQ/8.25kQ divider shown in figure 5.

The third type of scan is used for the 40 to 60 GHz band and uses the external mixer with the

counter (see figure 4). The measured frequency is corrected for the Gunn oscillator frequency to

give the actual optical beat frequency. In this case YAG3 is cooled and YAG2 is swept to

achieve sample spacing of about 5 MHz.

A fourth option is a phase-locked mode which steps to specific frequencies. It can synthesize

specific frequencies between 10 Hz and 1 GHz. Phase-locked operation is described in the next

section.

10
These scan options are accessible through a user interface program called HETSETUP.BAS.

This interface program is designed to be run as an executable which calls other programs, also in

compiled executable form, which run the dififerent scan options listed above. The measurement

number is recorded in a master file and is incremented with each scan, so that subsequent

measurements use a unique file number. A setup file for each calibration records the file numbers

that were used for that calibration along with the detector name and type and customer name. Data files generated by the programs described above contain the corresponding setup file name,

data file name, temperature, and relative humidity for the scaiL The files then split into four

columns that contain fi-equency (in gigahertz), rf power (in decibels with respect to 1 mW),

current (in amperes), and normalized response (in decibels).

3.2 Phase-locked operation

Measurements v^th resolution of about 100 kHz are desired at low fi"equencies but the minimum

practical resolution for swept measurements is greater than 30 kHz, giving the final data spacing

of about 300 kHz. A method that gives very fine resolution and solves problems related to scanning the beat fi"equency through dc is to synthesize the beat fi-equency between the two

lasers using a phase-locked loop. Early work demonstrating phase-locked operation of two

nonplanar Nd:YAG lasers operated at low frequencies and displayed small capture ranges. Later

work by several authors [12], [13] showed broadband operation above 10 MHz. However, most

high bandwidth phase-locking circuits have minimum operating frequencies of a few tens of

megahertz and are not suitable at the relatively low frequencies required for some applications.

We have solved this problem by generating a secondary optical signal whose frequency is ofifeet from the main signal by 70 MHz (see figure 2). The phase locking electronics then see a signal near 70 MHz when the ofl&et frequency of the main signal is very low.

The electronics for the phase-locked loop (PLL), shown in figure 6, were developed by the Time

and Frequency Division at NIST [14], [15]. The local oscUlator (LO) is offset from the rf

frequency by 15 MHz (see figure 5). The difference frequency between the LO and rf signals is sent to a phase comparator to compare with an inexpensive TTL oscillator operating at 55 MHz,

and generates an error signal that is filtered and fed back to the laser frequency control. For the

sign ofthe error signals used the beat frequency/2 at BS2 is offset from/I ( at BSl) by 70 MHz

and is offset from the LO by 15 MHz. For the main signal at BSl to have a frequency of 1 MHz,

the beat frequency at BS2 is 71 MHz and the LO is set at 16 MHz. The frequency stability is limited by the relative stability ofthe 55 MHz TTL oscillator, 70 MHz crystal oscillator, and

synthesized LO. We have not attempted to stabilize these oscillators relative to each other.

Phase-locked frequency control in our heterodyne system works well between about 300 kHz

and 1 GHz. Unfortunately, phase-locked operation below 100 kHz is unstable, probably due to coupling between the laser relaxation oscillations and the phase detection circuit.

11
LO

RF

to piezo and

thermal controls

Mixer

ECL

comparator

TTL crystal

oscillator 60 MHz

Low-pass

Filter f 55 MHz

ECL phase/frequency

detector

R2

C1 R3 —1 1—

R1

AAATTL to ECL

Low-pass

Filter

A Loop filter

Manual gain

adjust

Figure 6. Schematic of PLL box from figure 5. Component values for the loop filter are Rl=137

Q, R2= 158 kQ, R3= 3 Q, and Cl= 1 ^F.

3.3 Power matching

The NIST heterodyne system is designed to operate with 100 % modulation depth and to use the

normalization described by eq (6). Control ofthe modulation depth is achieved by three features

ofthe system: mode matching ofthe radiation incident on the detector from each laser,

polarization matching, and power matching. Mode matching is achieved by launching the laser

beams into roughly 2 m of single mode fiber. Polarization matching is provided by the polarizing

isolators. Any birefiingence in the optical path following the isolators changes the relative

polarization ofthe two laser signals by a negligible amount. The power from each laser is matched by fine adjustments ofvariable beam splitters VBSl and VBS2. Typically, a high power

is required for the high speed detector used for frequency measurement, so the ratio of the power

12
going to the frequency measurement eissembly to the power going to the specimen is controlled

by VBS3.

Effects which cause the modulation depth to vary from 100 % should be minimized. The optical

return loss oftest devices should be higher than 20 dB to reduce multipath interference in the

fiber delivering light to the detector. Fiber connectors used should mate with minimal gap to avoid wavelength dependence in the insertion loss. Since the power from YAG3 which is coupled into the fiber changes when it is thermally tuned, VBS2 must be used to adjust its power

before the second part ofa 40 GHz scan.

4. Uncertainty in combined photodiode/power sensor normalized response

Uncertainties in the measurement of a photodiode with a current monitor and a receiver without a current monitor are different. Uncertainties peculiar to a receiver without current monitor will

be discussed in Appendix D. Typical uncertainties for the photodiode are listed in table I. [16]

Type A uncertainties are uncertainties that can be calculated using purely statistical analysis and

include repeatability, offset correction, and reproducibility (environmental effects). Type B

imcertainties are inferred by other means. The coverage factor of 2 gives approximately 95 %

confidence that the true value ofthe frequency response lies within the interval [measured value

± expanded uncertainty] if all the uncertainties obey a Gaussian distribution. We anticipate that

most customers who are interested in calibrations of this type will be mostly interested in

uncertainties that might change the shape ofthe frequency response and are not as concerned

with the absolute response. The type B uncertainties that are important for this type oftransfer

standard are quantities which might vary with frequency or which might change during the

measurement. Range-scaling uncertainties in the meter are included in the uncertainty budget

since the range setting might change during the measurement. The current monitor is also

included because its offset may drift. Uncertainties in the 50 MHz calibration standard in the

power meter are not usually included, although they are outlined below. A standard that can not

be connected to the 50 MHz reference calibrator introduces an arbitrary shift (or scale factor in

linear units) that can change between measurements, and which must be corrected. Optical

connector repeatability is not an issue in the normalized response measurement since the

response is normalized to the dc photocurrent, which is proportional to the power actually hitting

the detector. We assume that any insertion loss in the connectors has negligible wavelength

dependence, rf power meter range scaling and bias current monitor uncertainties are the

dominant contributions to the uncertainty budget. The total uncertainty given in table 1 is the

square root ofthe sum of the squared uncertainties (rss). Details ofthe uncertainty budget are given below.

4.1 Type A uncertainties

Repeatability: Since the heterodyne system uses a swept measurement, different scans wiQ not

repeat exactly the same frequencies, nor will the measured frequencies exactly coincide with the

requested frequencies. Two scans are made in both forward and reverse directions, giving a total

13
Table 1. Summary oftypical frequency response measurement uncertainties for photodiode

with rf power sensor and 3 dB attenuator.

Source of uncertainty Uncertainty,

A. 1 9 / U

Type A

Measurement repeatability s <0.05

Offset correction 0.007

Environment effects Sgifm GHz) 0.106 + 0.00064/+

lype a

Meter range scaling 0.5

Meter offset 0.023

Optical power drift (power matching) 0.06

Bias current measurement 0.08

Power sensor noise, @ 0.05 GHz 0.025

@-3 dB point 0.05

Total uncertainty (rss), @ 0.05 GHz 0.52

@ -3 dB point, 31 GHz, 4 scans 0.63

Expanded uncertainty (coverage factor k = 2),@ 0.05 GHz 1.05 (0.05 dB)

@ -3 dB point, 31 GHz, 4 scans 1.25 (0.05 dB)

offour scans over each range. The response and its standard uncertainty at the requested

frequencies are found using the following method. Let M^(/) be the measured response at frequency/ The calculation is usually performed withM^(/) in dB with negligible error because

the scatter in the data is small. The response at the requested frequency jc is estimated by a

kernel-type smoother [17]. Specifically,

( A

(x)

; = 1 (9)

/ = 1

14
where b is the bandwidth parameter and K{ ) is the kernel function, which typically has the

following properties:

1. ^ 0 for all ^

2. rmdi = 1 (10)

J - 00

3. ^(-0 = A:(OforaU^.

That is, (x) is obtained as a weighted average of frequency responses around x. The kernel

function assigns the weight to each point, and the bandwidth b determines the size of the region

around x for which M^(y] ) receives relatively large weights. Frequently used kernels include

rectangle, triangle, and Gaussian functions. The rectangular kernel assigns equal weight to the

points inside the rectangle and ignores the points outside. Both the triangle and Gaussian kernel

assign the most weight to the points that are closer to x. In our applications, we use the

Gaussian kernel

-I' m = 1 e . (11)

0.37^271

The kernel is scaled to have 25th percentile of -0.25 and a 75th percentile of 0.25.

Many methods have been proposed for bandwidth selection. [18] In practice, b can be

determined by the average frequency increment in the scan, and the desired number ofpoints

used to calculate (jc) . For example, if the Gaussian kernel is used, the average frequency

increment is h, and roughly p points on each side ofx are to be included; that is, |x -

f.\ < ph.

Any point with |(x - f)/b\ > 1 receives almost 0 weight; that is, we want |(x - f)/b\ < 1 or b « ph. The "resolution" of the measurement is then about 0.37ph. We typically take data so

that the spacing between acquired points is 0.1 to 0.2 times the final data spacing (resolution)

required. We then use p = 2 giving an approximate resolution of 0.07 to 0. 14 times the

customer's requested frequency spacing.

The kernel-smoothing technique, like all other smoothing techniques, has an error associated

with it. This error is small when the response is roughly linear with frequency, but may become

significant in regions with large curvature. The approximate mean-squared error of the kernel

estimator (x) is given by [19]

15
MSEix) = 0.004 6S5 b^c^ + 0.762 42

^

where is the second derivative of R^Cy) evaluated aty = x using a spline method and is the

variance of the response of a given scan. This variance, however, is unknown and needs to be

estimated. A technique [20] that works well and can be used to obtain an estimate of is to

"detrend" the data locally and use the sample variance of the detrended data. The detrended

residual value at is defined as

r. = R2(/;.) - —lill^ (13)

and the variance estimator is found by

62 = —A ^ ^2

n-l

3(«-2) ^2

Let Rf (x) and MSE/x) be the estimated response and the mean-squared error at fi*equency x of

the rth scan. The estimated mean response based on m scans is

-2.,. _ f (15)

E m

and its standard error is

s{x) = ^

(16)

where

E(=1

(R; (x) - R"(x))'

w - 1

(17)

16
is the between-scan uncertainty and

£^ MSEfx)

i=\ m (18)

22.5 24.5

1 ' I

' r 23.0 23.5 24.0 Freauencv. GHz

Figure 7. Dots represent the measured response,

denotes the interpolated responses at requested frequencies based on a cubic

spline fit, and x denotes the Gaussian

kernel smoother.

is the uncertainty within a scan due to kernel

smoothing. Repeatability is a function of the

rf connector type, frequency, electrical

mismatch, and other factors that cannot be

controlled.

Other smoothing techniques can also be used,

in particular, spline methods. We did not use

a spline method to interpolate the

measurements because spline methods give

unsatisfactory results near the end points, use

arbitrary knot spacing, and give a poorly

defined resolution. In addition, the splines

over-smooth the peaks and valleys. For

example, the dots in figure 7 represent the

measured response, denotes the interpolated responses at requested frequencies based on a

cubic spline fit, and x denotes the interpolated responses at requested frequencies based on the

Gaussian kernel smoother. In most cases, both methods agree well, but clearly the spline method

gives poorer results in the neighborhood of23.5 GHz. Uncertainty due to the kernel smoothing

is negligible on most detectors that we have calibrated; for example, < 0.002 dB for the data in

figure 6.

Offset correction: When the power sensor gain is not scaled to the 50 MHz power reference (or

when the frequency response is normalized to the optical power instead of the photocurrent), a

correction due to the scale factor must be included. The scale factor (offset on a logarithmic

scale) is constant for each scan and does not have a frequency dependence if coherent multipath

effects are insignificant. The fiber connector insertion loss repeatability is typically about 0.1 dB

and is larger than other uncertainties in the measurement. Variation in the scale factor is normally about 0.005 dB. However, if the meter is used with a different power sensor and it is scaled relative to the 50 MHz reference, the scale factor could change by a few tenths of a

decibel. Also, scans over the low and high frequency ranges are often made separately and with

different data spacing, so a method for "splicing" the two overlapping frequency ranges is required. This is accomplished by shifting all the scans to a common level (on a logarithmic

scale) selected by a least-median technique. Using this technique, the individual scans are shifted

by averaging over all desired frequencies where scans overlap. The entire data set is then

normalized to give 0 dB at some frequency, typically 50 MHz. The absolute response then has a

17
relatively large uncertainty, while the shape is specified with much lower uncertainty. The

technique described below is valid for logarithmic data if the offsets or outliers are less than a few tenths ofa decibel; otherwise, the calculation should be performed with responsivity in linear

units. If this were the case, the procedure below would need to be modified to accommodate a

linear scale factor instead of a shift.

If there are m scans over «o overlapping frequencies/, then offset adjustment factors can be

calculated by minimizing the expression

E - a,, Ufi) - a,, •••

, Ufi) - «J ' (19)

/=i

where d( - ,

- , —, • ) is an "appropriate" distance metric and 'y^ = 101og(M^ ) or 101og(g^ ), as appropriate for the particular measurement (see Appendices A and D for definition ofg). A

possible metric for our application is absolute deviation from the mean. Let

Y(/;) = E^ (20)

be the mean response at / and define the distance metric as the sum ofthe distance (absolute

difference) between "ijif) ~ a. and the mean response Y(yj) m d{iw - - , yM) - O = E IyM) - - yifi)\ ' (21)

j=i

The expression we need to minimize is

m "o EEiW) - Y(y;)i • (22)

j=\ /=i

Given the sequence yi, jVj, ...,yk, the value ofa that minimizes

E \yr^\ (23)

j=i

isa = median of>'i, >'2, y/,. Thus the solution for a, is that a, is the median of

%(A) - Y(/,), ?//2) - Y(/2), •••

, ?//„„) - Y(/„„) . (24)

18
The ofifset uncertainty can be characterized by the standard deviation of the a, If>'„

independent Gaussian random variables, the variance ofthe median of>^,, is approximately equal to [21]

., y,, are

-Kvariy.)

(25)

2k

Now, by expanding y(/) into a sum, we can show that

m

vai (26)

so an estimate of the variance ofa, is given by

t-l)7id^

(27)

where is the variance estimate of scanj given in eq (14).

Environmental conditions: Another source ofuncertainty is due to environmental conditions

such as temperature, pressure, relative humidity, and other unknown interfering factors. A

check standard was constructed in an attempt to evaluate these effects and to track changes in

the measurement system. Unfortunately, each transfer standard is affected by these parameters

differently, and it is impossible to evaluate the long term reproduceability or stability of each

customer's transfer standard. We intend that our measurements of the check standard to be used

as a measure ofthe system reproducability and that the reproducability of the transfer standard

be left to the customer.

The check standard consists oftwo photodiodes, each with regulated bias and bias monitor

circuits. The photodiode labeled "Lowband" has a nominal bandwidth ofabout 16 GHz and is connected to a 3 dB attenuator, a 3.5 mm to N adapter, and a commercial rf power sensor

(diode) that operates between 100 kHz and 4.2 GHz. The photodiode labeled "Highband" has a nominal bandwidth ofabout 30 GHz and is connected to a 3 dB attenuator and a commercial rf

power sensor (diode) that operates between 50 MHz and 50 GHz. Both photodiode assemblies

are permanently mounted in a box with a power supply and access to the bias monitor and power

sensor. Measurements were made on both bands over a year's time. Because the photodiodes were permanently joined with the power sensor, no connection to the 50 MHz reference source on the power meter could be made. Hence all measurements were simply normalized to the

value at 50 MHz. The nominal response curves and deviation from the mean are shown in

figures 8 through 10.

19
00D

0)

0)

c

o

Q.

</) 0

21 -0.4 —

0.4

0

D

0)N

m -0.8 — OS

E

v_O

c

c

(00

-1.2 —

1.6

0

1

' r 20 40

Frequency, GHz

60

1 2 3

Frequency, GHz

Figure 8. Mean normalized response of the check standard shifted to give 0 dB at 0.05 GHz.

Upper graph is highband, and lower graph is lowband.

20
^ 0.08

f-0.04

Q-0.06

C^0.04

0

T

20 40

Frequency, GHz

60

60

1

' 1

20 40

Frequency, GHz

Figure 9, Measurements on highband of the check standard. Top graph shows deviation from the

mean which neglects measurements on July 25, 1997. Bottom graph shows the 2o

variation with a fit to a quadratic equation. Spikes at 52 GHz go out of the range of the graphs.

21
0.04

c

(00

E

o

c

q

CO >

0D

0

D)

(Q

I

1

'

\ P^requency, GHz^

£ 0.01

0.006 —

0.002 —

-0.002 —

-0.006

August 4, 1997

vember 10, 1998

j March 3, 1998

January 1^, 1998

October 27 . 1 998 December 2, 1997

April-June, 1996

18 20 22.

Temperature, deg. F

24

Figure 10. Top graph shows the deviation from mean of measurements on the check standard

lowband. Influence of measurements made 8/4/97, 4-6/98, and 1 1/10/98 is clearly seen. When these data sets are dropped, 2o is about 0.05 dB. Bottom graph shows

correlation between temperature and average offset from mean of data sets. Error

bars are one standard deviation of the data and do not include inaccuracy of the

thermometer.

22
Measurements on the highband photodiode consisted of swept scans over the following

approximate ranges: 50 MHz to 1 GHz, 500 MHz to 40 GHz, and 38 GHz to 55 GHz. Two

scans were made in both forward and reverse directions, giving a total of four scans in each

range. The room temperature and relative humidity were measured before each scan. Where

scans overlap in frequency, the lower frequency scan was always used in the final analysis. A

plot of the deviation from the mean (see figure 9) shows monotonically increasing uncertainty

with frequency with large spikes near 43 and 52 GHz where the frequency response of the

standard has notches. There is not a jump in the deviation in the region of 40 GHz, where

frequency is measured with the Gunn oscillator and mixer, showing that frequency accuracy is not a major problem in the 40 to 55 GHz region, although this may be part of the reason for

large uncertainties at the notches at 43 and 52 GHz. The measurement performed on July 25,

1997, shows a significantly higher deviation than the other scans below 23 GHz, but then is in

the middle at higher fequencies. If this data set is neglected, the la deviation from the mean fits roughly a second-order polynomial in frequency,

s,{f) = «o ^i/ ^ ^2/' • (28)

Measurements on the lowband photodiode consist of a swept scan over the range of a few

megahertz to 4.2 GHz and a set of data acquired using the phase-locked loop between 300 kHz

and 600 MHz. Swept measurements were made in the forward and reverse directions giving a

total of four scans and three sets of data were acquired using the phase-locked loop. A plot of

the deviation from the mean (see figure 10) shows that measurements taken on August 28, 1997,

fall above the mean and measurements in April/June 1998 fall below the mean. Both data sets have a large ripple indicating that the ripple in the frequency response was distorted in these

measurements. In an attempt to find a correlation between room temperature and these

deviations, the average temperature was plotted against the average deviation for each data set showing a possible trend. A final measurement taken November 10, 1998, with the room

temperature intentionally lowered appears to confirm the trend. Whether this trend is the cause ofthe slightly higher value in the data below 23 GHz in the highband data is unkown.

Since the lowband data for August 28, 1997, and April/June 1998 were shown above to be

outliers, they can be discarded. Then the data from the lowband measurements and highband

measurements are consistent and the expression for the environmental random variation in eq

(28) is still valid if the laboratory temperature stays at 21 .55 ± 0.28 °C. It is possible that

environmental uncertainty is mostly due to environmental effects on the check standard and not

on the measurement system. However the check standard is the most stable device that we have

been able to construct, and its stability cannot be measured independently from the stability of

the measurement system. Until a better check standard is available, the reproducability of the

check standard measurements will be used as a measure of the system stability. This is a good

assumption since the contribution of is not really significant below about 40 GHz.

The combined type A uncertainty is then found by adding the components of uncertainty in

quadrature [14]. That is, we use

23
2 m

(29)

as the total type A uncertainty in Table 1, is obtained using the offset adjusted data in eq (16),

m is the number of scans within the particular frequency band, and is the maximum of the ^

(see eq (27)) for all the scans and frequency bands. If offsets are not required 5^ = 0.

^

4.2 Type B uncertainties

Meter range scaling: This uncertainty is due to the inaccuracy of the gain for different power

ranges. The absolute accuracy of the power meter is specified by the manufacturer as 0.5 % and

is assumed to be lo. This is conservative; if the specified uncertainty were actually 2o the

estimated calibration uncertainty would be reduced. Also, the uncertainty is assumed to obey a Gaussian distribution instead of rectangular distribution. This is also a conservative assumption.

The meter range scaling uncertainty is the dominant contribution to the uncertainty budget in

Table 1.

Meter offset: The settability of the zero level, specified as one least significant count on the

display or 0.001 dB (0.023 %), is assumed to be lo. Additional zero setting error is 0.5 % of

fiill scale deviation of lowest range. For the diode power sensor this is 5 pW.

Opticalpower drift (power matching): Power from one or both ofthe lasers that is coupled into

the fiber may drift during the measurement. Also because of the long coherence length ofthe

Nd:YAG laser, multipath effects cause the power matching between the two lasers to vary with

frequency. The power from each laser does not drift by more than 4 %, giving a 0.04 %

uncertainty in the ratio of eq (6). This quantity isassumed to be lo for one ofthe lasers. Adding

the uncertainty in quadrature for two lasers gives ±0.06 % uncertainty.

Bias current measurement: The scale factor in the current monitor used in the check standard

has a temperature coefficient of about 0. 14 %/°C. Using the laboratory drift of about ±0.28 °C,

the uncertainty in the bias current is 0.04 %. Squaring gives a 0.08 % uncertainty in the

normalized response. If a bias current monitor is not provided, the voltage drop across a 1 kQ

resistor in series with the detector is measured. The temperature coefficient ofthe resistor is 0.01 %/°C giving 0.0028 % uncertainty. The specified accuracy of the voltmeter on the 300 mV

range is 0.0025 %, which is assumed to be lo. Other current monitor circuits may have different

uncertainties. The dark current and/or offset of the monitor circuit is subtracted from the

measured current.

Power sensor noise: This is the uncertainty associated with the background noise ofthe

detector, wliich is about -75 dB (re: 1 mW) for the diode power sensor. Using 100 \iA total

24
bias current and a 3 dB pad gives an error of 0.013 % at low frequencies. The background noise

is about -40 dB (re 1 mW) for thermal power sensors.

Reference power uncertainty: If the transfer standard is of the type where the power sensor is to be disconnected to scale the power meter gain, an uncertainty from the 50 MHz reference source needs to be included, since it assumed that the iatent ofthe sensor is to calculate absolute

modulation depth. The power reference accuracy is specified by the manufacturer as ±0.9 %

RSS for one year and is assumed to be lo. Mismatch between the power sensor and the

reference gives an additional error. The reference is specified as having a maximum VSWR of

1.05, giving p <. 0.024. The reflection coefficient of the power sensor at 50 MHz typically varies

between 0.015 and 0.03. An upper bounds for the mismatch error is |1 ± (0.024)(0.03)p - 1 =

±0.0014, or ±0.14 %. Modeling the distribution as uniform, gives a standard deviation of 0.08

%.

Gunn oscillatorfrequency drift: Uncertainty in the normalized response due to frequency drift

is given to first order by

6(101og[m2(/)]) = bfslope. (30)

A maximum slope is about 3 dB/GHz, the drift of the oscillator is about 9 MHz/°C and the

laboratory temperature drift is about ±0.28 °C worst-case. This gives an uncertainty in the

normalized response of about 0.008 dB or 0.17 %.

Hysteresis: Averaging internal to the power meter can cause the meter response to effectively

shift and smooth sharp features in the frequency response. This is avoided by using 0.01 dB

resolution, with the default averaging factor instead of 0.001 dB resolution and making two

scans in both forward and reverse directions. The result of scanning in opposite directions is to

(a) average errors and (b) flag regions where there are problems by showing large InterScan

uncertainty.

Relative intensity noise: Laser relative intensity noise (RIN) could degrade the accuracy of the

heterodyne measurement system by adding a small amount of power to the measurement from

both the spontaneous-spontaneous beat note and the signal-spontaneous beat note. In the

Nd:YAG system, the signal-spontaneous beat noise causes side lobes are >30dB (re: carrier)

below the carrier at about 100 kHz offset. This places a small error on the resolution of the

system, but this error is insignificant for most real detectors. The spontaneous-spontaneous beat

noise is many orders of magnitude lower and is an insignificant source of error.

25
5. Quality Control

5.1 Calibration notebook

A notebook which describes the measurement setup for all calibrations, voltage range range used

for each scan and corresponding frequency range, biasing conditions, adapters, and any other

pertinent information will be kept.

5.2 Checks on each calibration

A system failure which would possibly give erroneous results occurs when the lasers are not truly

operating in a single mode. This is the usual problem in laser temperature regions where the

laser mode is unstable and near a mode hop. A spurious mode has also been observed in YAG3

for certain tuning conditions. Precautions which will be taken to ensure rejection of invalid data

due to spurious modes include checking the optical spectrum analyzer at the beginning and

occasionally during a scan.

Frequency scans over adjacent bands should overlap both in frequency and in magnitude.

Typical offset between scans is less than 0.02 dB. If the bands do not overlap, and the power

meter scale factor was not set with a different power sensor, then the operator should search for

the cause of the problem.

The bias current for each scan wUl be checked and should change by no more than 5.6 %.

5.3 Periodic checks

Power stability: When the data for a scan are being acquired, the total photocurrent is monitored and during the analysis of the data it is checked to make sure it did not vary out ofthe

specified 5.6 % (lo) total allowed deviation. However, periodically, the photocurrent ofthe

each laser will be monitored during a frequency scan to check for problems.

Gunn oscillatorfrequency: The frequency of the Gunn oscillator used for downconverting

frequencies in the 40 to 60 GHz band is subject to change with temperature and external stresses,

such as from mounting clamps. The frequency and temperature dependence ofthe Gunn

oscillator will be checked periodically or after major changes to its mount. This is accomplished

by mixing the oscillator's signal with a source of known frequency near 50 GHz and measuring

the difference frequency.

Check standards: A check standard is used to verify the operation ofthe heterodyne system A

fixed reference standard has been implemented (designated NIST ET) and is described in the

section on environment sources ofuncertainty above. Periodic measurements on this standard

wUl be performed to verify the system integrity and to obtain better information on the check

26
standard itself. As further measurements on the check standard are acquired, the estimate of

will be modified.

5.4 Future system changes

Possible future system changes could include, but are not limited to, replacement of YAG2,

modification of YAG3, use of an electrical spectrum analyzer for fi-equency measurement, use of

an additional laser for extending the fi^equency range above 60 GHz, replacement of the 40 GHz

bandwidth detector, or corrections for system "bugs." Future system changes will be

documented in a calibrations notebook which will be kept with the system. Instructions which

are specific to the NIST heterodyne system and data analysis are also kept with the system

5.5 Transfer standards which are suitable for calibration

The transfer standards must include a connectorized single-mode input fiber. The fiber

connector should be clean and scratch fi-ee. The photodiode response should be fi-ee of sharp

features to minimize calibration uncertainty. The photodiode must have either direct access to the bias point or have an internal current monitor circuit which is stable so that correct

normalization can be achieved during each measurement scan. Coaxial rf connectors must be of

a precision type, fi'ee of debris and in good condition. The power sensor must mate with a power meter which can be controlled via a standard GPIB interface. Depending on the make and

model ofthe power meter used, the customer may be asked to include the power meter with the

transfer standard for calibration. Transfer standards which are determined by NIST to not meet

these requirements will be returned at the customers expense and will be charged for any labor

which may have been required at NIST to determine the suitability ofthe transfer standard.

6. References

1 . Optical interfacesfor equipments and systems relating to the synchronous digital hierarchy.

CCITT (now ITU-TS) recommendation G.957, Geneva; 1990.

2. In this work, all ratios are stated in decibels electrical, that is, 201og(A2 /A,) where A is a

voltage or current. Absolute powers are stated with the reference level, such as - 10 dB (re: 1 mW) instead of - 10 dBm as suggested in Letter symbols to be used in electrical technology,

Part 3: Logarithmic quantities and units, lEC 27-3, Geneva; 1989.

3. Valdmanis, J. A.; Rudd, J. V. High-speed optical signals demand quick response. Laser

Focus World, pp. 141-147; Mar. 1995.

4. Humphreys, D. A.; Lynch, T.; Wake, D.; Parker, D.; Park, C. A.; Kawanishi, S.; McClendon,

M.; Hemday, P.; ScWafer, J.; Gnauck, A. H.; Raybon, G.; Hawkins, R. T.; Jones, M. D.; Goll,

J. H. Summary of results fi-om an international high-speed photodiode bandwidth

measurement intercomparison. Digest, Optical Fibre Measurements Conference, York,

U.K., pp. 69-72; 1991.

27
5. Gifford, A. D.; Humphreys, D. A.; Hale, P. D. Comparison ofphotodiode frequency

response measurements to 40 GHz between NPL and NIST. Electron. Lett., vol. 3 1, pp.

397-398; 1995.

6. Hale, P. D.; Franzen, D. L. Accurate characterization of high speed photodetectors. Proc.

SPIE, vol. 2022, pp. 218-227; 1993.

7. Bowers, J. E.; Burrus, C. A. Ultrawide-band long-wavelength p-i-n photodetectors. J. Lightwave TechnoL, vol. 5, pp. 1339-1350; 1987.

8. Wangsness, R. K. Electromagnetic Fields. New York: John Wiley & Sons; pp. 440-443;

1979.

9. Fundamentals ofrfand microwave power measurements, Hewlett-Packard Application Note

64-lA; 1997.

10. Wong, B.; Kosiorska, H. Factory measurement solutions to predictable field performance

of single mode fiber optic connectors. Day, G. W.; Franzen, D. L eds. Technical digestSymposium on optical fiber measurements; 1992. Natl. Inst. Stand. (U. S.) Techno1. Spec.

Publ. 839; pp. 23-28; 1992.

11. Gallawa, R. L.; Li, X. Calibration of optical fiber power meters: the efifect of

connectors. Appl. Opt, vol. 26, pp. 1170-1174; 1987.

12. Kazovsky, L. G.; Atlas, D. A. A 1320-nm experimental optical phase-locked loop:

performance investigation and PSK homodyne experiments at 140 Mb/s and 2 Gb/s. J. Lightwave Technol. vol. 8, pp. 1414-1425; 1990.

13. Williams, K. J.; Goldberg, L.; Esman, R. D.; Daagenais, M.; Weller, J. F. 6-34 GHz offset

phase-locking ofNd:YAG 1319 nonplanar ring lasers. Electron. Lett. vol. 25, pp. 1242-

1243; 1989.

14. D'Evelyn, L.; Hollberg, L.; PoPovic, Z. B. A CPW phase-locked loop for diode-laser

stabilization. IEEE MTT-S Int. Symposium Digest, San Diego, pp. 65-68; May 1994.

15. D'Evelyn, L. Noise characterization ofextended-cavity diode lasers using a newphaselocked loopfor diode laser stabilization. M.S. Dissertation, University of Colorado,

Electrical and Computer Engineering Dept.; 1994.

16. Guide to the expression ofuncertainty in measurement. ISO, Geneva; 1993.

17. Silverman, B. W. Density estimationfor statistics and data analysis. London: Chapman and

Hall, pp. 34-72; 1986.

28
18. Hadle, W. Smoothing techniques with implementation in statistics. New York: SpringerVerlag, 1991.

19. Altman, N. S. An introduction to kernel and nearest-neighbor nonparametric regression. Am.

Statistician, vol. 46, pp. 175-185; 1992.

20. Rice, J. Bandwidth choice for nonparametric regression. Ann. Statistics, vol. 12, pp. 1215-

1230; 1984.

21. Mood, A. M.; Graybill, F. A.; Boes, D. C. Intoduction to the theory ofstatistics. New York:

McGraw-Hill; p. 257; 1974.

29
Appendix A: Use of transfer standard in a ratio system

A.l Application of heterodyne formalism to arbitrary modulation depth

Signals that do not have 100 % modulation depth are commonly used in optoelectronic test equipment. An arbitrary modulation depth can be synthesized by varying one or both of the laser

powers in the heterodyne system and can be modeled using the same formalism as section 2.1

above. The resulting equations are applicable to any source with an arbitrary modulation depth,

such as a laser with direct or external modulation. A notationaUy simple way to change the model

is to let Pa^g denote the total average optical power incident on the detector, Pqi = ccP^^ and P02 = (\ - ccJPavg' Then the total optical signal incident on the photodiode is

A0= Pavg 2P - a) cos(27t:/0 . (Al)

The average optical power is then P^g and the absolute modulation depth is 4P^g[a(l - a)p , or,

in fractional units, the fractional modulation depth Mq is

^0 = ^""p

- 2Va(l - a) , (A2)

where P^^x ^mm ^re the maximum £ind minimum optical powers incident on the receiver. The

mean squared currents in the photodiode are

<'f/>=V*'(0) (A3)

and

<i,f'>=2P,,^M^ - CL)R\(^) , (A4)

so the ratio of the powers is 2P

Hence, the modulation depth of an arbitrary source can be measured with a detector ofknown

normalized response. The combined normalized response can be used in eq (A5) ifP„ is used

instead of P^. If the normalized response is only known to within a constant scale factor, the

modulation depth is known only within a scale factor. This is the case if the photodetector and rf

30
power sensor are permanently attached and the combined normalized response is further

normalized to give 0 dB at 50 MHz.

In the case ofthe receiver without access to bias current (see Appendix D), the modulation depth

ofan unknown source can be found in terms ofthe measured rf power, average optical power,

and known voltage frequency response as

2 "^^L^rf - -—T- • (A6)

avg

A.2 Calibration transfer using ratio of measurements

The fundamental problem in making photodetector frequency response measurements is getting

accurate knowledge of the modulation depth of the source. The modulation depth of the

heterodyne beat signal is known from fundamental principles, but the modulation depth of a

directly modulated laser diode is not. The modulation depth of a Mach-Zehnder modulator can be

measured, but careful control of bias and net optical transmission are required for accurate

measurements. A transfer standard calibrated using the heterodyne techniques described above

and the ratio measurement system shown in figure Al can be used to obtain accurate knowledge

of the modulation depth of these sources. The ratio system consists of a modulated light source with unknown modulation depth and a 1 x 2 coupler. The coupling ratio does not need to be

specified. The signal from the coupler is delivered to a DUT and to a reference photodiode

attached to a rf power sensor. The modulation depth of the source can be calculated using eq

(35) and the known frequency response of the transfer standard photodiode. Then the frequency

response of the DUT is

p

(A7)

Calculation of the DUT's frequency response this way includes calibration uncertainty of both the

power sensors used to measure from the standard detector at NIST and at the customer's

laboratory, giving a total expanded uncertainty of 0.2 dB or more for the modulation transfer

fimction. Combined with other fectory uncertainties, this may give an imacceptable uncertainty

for the intended test system.

One possible alternative is to calibrate the response of the photodiode combined with the rf power

sensor. This method totally eliminates uncertainties due to power sensor calibration and

impedance mismatch. When the photodiode/power sensor combination transfer standard is used

in the ratio test system, the combined frequency response is used to find the modulation depth of

the source

31
The normalized response ofthe DUT is then found using eq (A5), the do photocurrent, and the rf

power (including aU calibration factors) from the DUT

In some test applications it may be preferable to measure the DUT's frequency response in terms

ofthe coupling ratio instead ofthe bias current or average power. This may be the case when the

DUT has poor dc stability or when the frequency response will be normalized to the response at a

specific frequency (eliminating the coupling ratio). In this case eqs (A8) and (A9) can be

combined to give

4t/7</) = ^^M^(/) p^ (Aio)

m:ref

where P is the ratio ofthe optical power coupled to the reference arm to the power coupled to the

test arm.

In the case ofthe amplified receiver described in Appendix D the combined response ofthe DUT

is

G'if)

2R,P^ (All)

L m

^Prf.DUT

,2 . . ——-:r • (A9) K 0-5 <idc:DUT> Rl

and can be used to find the modulation depth of the source from the equation

2RrPL I

The DUT's response is then

»>f2 _ L m:ref - • (A12)

32
^^^^ = -JTi • (A13)

"avg-.DUT

A.3 Sources of uncertainty in ratio measurement system

The following is a brief list oferror sources that must be considered when designing a ratio test system.

Type ofphotoreceiver specimen: Equations (A9), (AlO), and (A13) include the rf power

measured from the DUT. If the DUT is a photoreceiver, the power must be measured with some

kind of electrical transducer that will indicate how much electrical power is being generated. The

calibration factor and impedance mismatch of this transducer must also be included in the

customer's error budget. However, if the DUT is a device such as a lightwave communications

analyzer or lightwave spectrum analyzer, which includes the transducer, the frequency response of

the device as a whole is measured, eliminating calibration factor and mismatch uncertainties.

Spurious harmonics: Nonlinearities in directly modulated lasers, over driving, or inaccurate bias

ofexternal modulators can cause some harmonic content [Al]. The rf signal generator can also

be a source of spurious harmonic and nonharmonic signals. Since the rf power sensor is not

frequency selective, energy at spurious frequencies gives a systematic error. Assuming that the

power sensor acts as a true rms detector, the powers add arithmetically. To keep the increase in

uncertainty below 10 %, the total harmonic power must be more than 22 dB lower than the

ftmdamental.

Relative intensity noise: Again, energy at frequencies other than the modulation frequency can

give systematic errors.

Wavelength dependence: Wavelength dependence ofa photodiode is a complicated ftinction of

the depletion region thickness, intrinsic layer thickness, bias voltage, and other parameters. The

variation in response due to wavelength can be as much as several tenths ofa decibel [Al]. Etalon

effects inside the detector can cause relatively large changes in the dc responsivity in a way which

is not directly measured if the bias current is not monitored. This effect can be minimized by

operating the ratio test system at the same wavelength as the original calibration or corrections can be performed if the transfer standard can be accurately modeled.

Reference

Al. Hale, P. D.; Humphreys, D. A.; Gifford, A. D. Photodetector frequency response

measurements at NIST, US, and NPL, UK: preliminary results of a standards laboratory

comparison. Proc. SPIE, vol. 2149, pp. 345-355; 1994.

33
Modulated

source

rf signal

generator

Fiber

coupler

Transfer

standard

rf power

meter

Figure Al. Ratio measurement system

34
Appendix B: Heterodyne system with unequal powers

The powers from each laser incident on the detector may not be equal. In this case it is possible

to formulate the expression for the normalized response in a way that eliminates error from the

assumption that the powers are equal used in eq (6). But first it is illustrative to calculate the

error induced by small deviations from equality. Let the optical powers be nearly equal, so that

Pqi = Pq and Pq2 = Pq+S, where 6 « Pq. Then

(Poi ^ Po2f = 4(i>^ ^ P^b) ^ b\ (Bl)

~2^Poi - P02f

(B2)

The last two lines ofeq (6) then follow from eqs (Bl) and (B2). The error caused by a small

perturbation can then be approximated as

2P_

2(PMP'(f)

0.5(2P^ + 6)

1 . A

Po

i

1 + IPo

(B3)

1-1

4

_6

Po)

35
Hence, a 4 % fluctuation in Pq2 will cause a

0.04 % or 0.0017 dB change in the measured

value of

On the other hand, large differences between

the laser powers can lead to large errors in eq

(6), even if the individual photocurrents are known. This error is shovm in figure Bl

This problem can be overcome by monitoring

the individual photocurrents and using the

exact ratio

rf (B4)

1—I—I—

I—I—I—

-25 0

1005/Po, %

Figure Bl. Error in ratio if total current is used

instead of product of individual

currents in eq (6).

Appendix C: The meaning of normalized response and 0 dB

A simple interpretation ofthe normalized response is the ratio ofthe rf responsivity (refered to the

output (rf) port) to the dc responsivity (refered to the bias (dc) port). A normalized response of 0

dB then can be interpreted as the dc and rf responsivities (as defined in eq (4)) being equal. Since

the rf responsivity is refered to the output port, the effect of any attenuators in the output path

will be shown as an offset. For example, a detector followed by a 6 dB matching pad will show a normalized response that approaches -6 dB at low fi"equencies.

Another interpretation for 0 dB is that the rf signal fi-om the photodiode is the maximum possible

without driving it into forward conduction. That is, if the dc current is ig then the rf current has a maximum of +2io and a minimum of 0.

The normalized response may be greater than 0 dB in steady state operation when the detector

has some gain or the detector includes some inductive resonance. In the latter case, the diode

current source drives an LRC tank circuit.

36
Appendix D: Photoreceiver with no access to bias monitor

D.l Expression of absolute responsivity in terms of measured quantities

Some optical receivers will not allow access to the dc photocurrent or bias current. In this case it

is neccessary to measure the rf power generated relative to the optical power incident on the

receiver. Since most high speed receivers are fiber-pigtailed and connectorized, it is difficult to measure the exact optical power that is incident on the receiver because of poor connector

repeatability. This Appendix considers this situation. At this time, measurement of a receiver that

does not have access to the bias current is offered only as a Special Test. It is desirable in some

applications to know the absolute voltage response of the photoreceiver. An amplified

photoreceiver commonly requires this type of characterization. This Appendix will demonstrate

the formalism used in calculating voltage response.

For a photoreceiver without a bias current monitor, the system in figure 1 includes a 10 dB

(nominal) optical coupler that monitors the total power fi*om the lasers during a fi'equency scan. The power incident on the DUT is inferred firom the coupler monitor port. The voltage that the

receiver delivers to a load Ri is given by

Vpi m = (Poj ^ ^02)^(0) ^ 2;?^G(/)cos(27c/0 , (Dl)

where G{f) is the amplified response ofthe detector (in YfW) at fi'equency/ The first term on the

right side is the average dc signal, and the second term is the rf signal. The voltage response is found in a similar manner to eq (6) and is

G\f) - , (D2)

avg

where P^^ = Pq^ + Pq2 is the average (dc) optical power incident on the detector and P^is the

electrical power delivered to a load Ri (the power sensor). This ratio is insensitive to small

changes in power fi*om either laser because the powers fi-om the two lasers are nearly equal. If

the voltage response is measured in dB (re 1 V/W), rf power in decibels (re: 1 mW), the average

optical power is in decibels (re: 1 mW), and R^ = 50 Q, eq (6) can be written as [2]

201og(G(/)) = 50 + - 29^^^ , (D3)

where 9 denotes a power measured in decibels (re: 1 mW).

D.2 Uncertainties of amplified receiver/power sensor combined response

Typical uncertainties for the receiver are listed in table Dl . The dominant uncertainties are fiber

connector insertion loss, power meter scaling, coupling ratio measurement, and power sensor

37
noise. Offset correction for the fiber connector insertion loss must be included to reduce the

relative uncertainty to an acceptable level. The uncertainty in the absolute level still remains and

can not be characterized within the tolerances that are typically required.

Coupling ratio: The coupling ratio is measured at the beginning ofa measurement and then the

10 % port is monitored during the scan to infer the power reaching the receiver. The fiber-to-air

launch fi'om the coupler monitor port into the power meter is terminated with a connector angled

at 8° to minimize coherent multiple path effects. Coherent effects due to nonzero return loss

(typically 30 dB) give repeatable systematic uncertainties of about 0.35 % (lo). Coupling ratio

drift during the scan is random and is included in the repeatability described above. An additonal

error in monitoring the optical power is sometimes observed if there are etalon effects in the thin

photodiode or between the fiber and detector. This causes the optical power measured by the

monitor to be different fi-om the actual power which is coupled into the photodiode and can be a

significant error over a large bandwidth.

Receiver linearity: Since the power detection is not fi-equency selective, harmonics ofthe

modulated signal due to nonlinearity give an error in the rf power measurement. To characterize

the receiver nonlinearity the optical power can be varied and harmonics measured on an electrical

spectrum analyzer. An optimum average optical power input is a compromise between receiver

nonlinearity and power sensor noise.

Receiver stability: Measurements at fi*equencies well above the amplified receiver's cut off can show additional instability. This may be related to the low open loop gain ofthe receiver

amplifier at these fi"equencies (see repeatability at 1500 MHz in table Dl).

Receiver noise: Some broadband receivers generate noise power significantly higher than the

power sensor noise floor, giving poor dynamic range in the heterodyne measurements. Often the

noise power is stable and can be subtracted fi*om the measured power in the heterodyne system,

giving significant improvements in the measurement dynamic range.

38
Table Dl. Summary of typical frequency response measurement uncertainties for the amplified

photoreceiver (nominal bandwidth of 0.8 GHz) with thermal rf power sensor.

Source of uncertainty Uncertainty,

k= 1,%

Type A

Measurement repeatability @ 0.010 GHz 0.05

@ 1.5 GHz 1.1

Oflfeet correction s^, 0.007

Environmental effects s, (/in GHz) 0. 106 + 0.00064/+

0.00024/2

Type B

Meter scaling 0.5

Meter offset 0.023

Coupling ratio 0.35

Optical power drift (power matching) 0.06

Receiver linearity, estimated from proto 900 0.1

Power sensor noise, @ 0.01 GHz 0.2

@ 1.5 GHz 3.8

Total uncertainty, @ 0.01 GHz 0.66

@ 1.5 GHz 4.0

Expanded uncertainty (coverage factor A: = 2), @ 0.01 GHz 1.3 (0.06 dB)

@ 1.5 GHz 8.0 (0.34 dB)

39
Appendix E. Sample calibration certificate

U.S. DEPARTMENT OF COMMERCE

NATIONAL INSTITUTE OF STANDARDS AND TECHNOLOGY

ELECTRONICS AND ELECTRICAL ENGINEERING LABORATORY

Boulder, Colorado 80303

REPORT OF CALIBRATION

Reference Receiver

XXX photodiode, Serial No. xxxxxxx XXX power sensor. Serial No. xxxxxxxxx XXX 3 dB attenuator, Serial No. xxxxxx

Submitted by:

Manufacturer's Name

Street Address

City, State Zip

The combined normalized frequency response ofthe Reference Receiver was measured at 1 .3 19

jim. The Reference Receiver consists of a photodiode wdth bias current monitor circuit, a 3 dB

attenuator or bias tee, and an rf power sensor. Measurements were made in the following

frequency bands:

1 . 200 - 900 kHz, 1 00 kHz steps

2. 1-40 MHz, 0.5 MHz steps

3. 41-1 19 MHz, 1 MHz steps

4. 120 - 600 MHz, 3 MHz steps

5. 630 - 4200 MHz, 1 5 MHz steps

6. 4250 - 40000 MHz, 50 MHz steps

7. 40100 - 50000 MHz, 100 MHz steps

Since the power sensors are fixed they can not be refered to the 50 MHz reference oscillator on

the power meter. Hence the "combined normalized response" M^(/) was further normalized to 1 (0 dB) at 50 MHz. Bands were spliced together with offset corrections typically on the order of

0.01 dB or less. The typical uncertainty in the measurement results is less than ±0.05 dB

(coverage factor k = 2) below 36 GHz and is dominated by uncertainties in the power meter.

Page: Page x of y

Calibration No.: xxxxx

Date of Report: month xx, 199x 40
Temperature control Desk top |-*-

computer Data acquisition

Fast photodiode

Intensity /\

control loop

D L

I

YAG#1

Intensity control loop

BS

D

50/50

BS

YAG#2

BS M

20 dB

RF

counter

Bias supply with

monitor

Fast photodiode

DUT

RF

power

sensor

Figure 1. Sinq)lified schematic ofthe NIST Nd:YAG heterodyne system. Labeled components

are beamsplitter (BS), mirror (M), isolator (I), lens (L), fiber (F), large area detector (D), and

integrating amplifier (A).

Above 40 GHz the noise ofthe power sensor becomes the dominant uncertainty.

Results for this Calibration are reported in a graph on page 8 ofthis report and on a 3.5 inch DOS

compatible disk in ASCII format.

NIST Nd:YAG heterodyne measurement at 1.319 fim

NIST uses a Nd:YAG heterodyne system for measuring scalar fi-equency response because the

excitation ofthe detector can be calculated fi'om first principles. A schematic of the heterodyne

system is shown in Figure 1 . The system uses two single-mode monolithic-ring Nd:YAG lasers

operating at 1 .3 1 9 \xm. The fi-equency ofeach laser can be tuned thermally to give beat

fi-equencies between dc and 65 GHz; the beats have a bandwidth ofthe order of 10 kHz. The

beat fi-equencies below 40 GHz are measured with an rf counter. Frequencies above 40 GHz are down converted to a fi*equency that the counter can measure by mixing with a Gunn oscillator

operating at 62.14 ± 0.015 GHz. As the frequency is scanned, data are acquired automatically.

The resolution ofthe system is limited by the scan rate, the frequency jitter, and the time constants

Page: Page x ofy

Calibration No.: xxxxx

Date of Report: month xx, 1 99x 41
ofthe data acquisition equipment. At present, the highest resolution achievable in swept

operation is about 300 kHz. Finer resolution (< 100 Hz) is attained using a phase-locked loop

(not shown in figure 1) at fi'equencies between 300 kHz and 1 GHz, The lasers are intensity

stabilized so that nearly equal power fi-om each laser is delivered to the photodiode as measured

by the bias current monitor. The combined laser beams pass through a polarizing isolator so that

the signal fi-om each laser is in the same polarization state when it reaches the photodiode

ensuring nearly 100 % modulation depth. The total optical power incident on the photodiode is

Ptotaff) = (^07 P02) ^ cos(27r/0 , (1)

and the photocurrent generated is given by

ip{f.t) = (Pqi + ^02)^(0) + 2y/P^7^(/)cos(2Tc/0 , (2)

where Pq] and Pq2 are the optical powers delivered to the detector from the lasers and R(f) is the

responsivity ofthe detector (in AAV) at frequency/. The first term on the right side is the dc

photocurrent, which flows through the bias supply, and the second term is the rf photocurrent

which flows through the rf load through a high-pass filter. The mean-squared photocurrent

generated by the photodiode is

where the first term on the right side is a dc term and the second is an ac term; hence

p dc 'rf < C > = < d > + < C > • (4)

These terms are proportional to the electrical power generated by the photodiode. IfPqi is nearly

equal to P02, then 2{PoiPo2) is, to first order, equal to O.SiPoi+Poif- The normalized frequency

response ^\f), which is defined as R\f)/R\0), can then be found by taking the ratio ofthe rf

electrical power to half the dc electrical power delivered to a load Rj^,

<iL>R. 0.5<ii>R

(5)

2{P^,P^,)R\f)R,

O.SiP^j ^ Po,fR\0)R,

.2 . ^ (/)

P^is a fimction of frequency. It includes corrections for sensor calibration factor and electrical

Page: Page x ofy

Calibration No.: xxxxx

Date of Report: month xx, 199x 42
mismatch, and is the power that would be delivered to an ideal load R^. Measurements are

typically made with transmission lines and loeids with characteristic impedance of 50 Q; however

these equations also apply to measurements with 75 Q systems.

Calculation of photoreceiver's frequency response in a ratio measurement system using the

normalized response, Bt\f) would include calibration uncertainty ofboth the power sensors at NIST and the customers laboratory, giving a total 2o uncertainty of about ±0.2 dB. Combined

with other uncertainties, this may give an unacceptable uncertainty for the intended test system.

Instead, this calibration measures the response ofthe photodiode combined with the rf power

sensor eliminating uncertainties due to power sensor calibration and impedance mismatch

correction. The combined normalized frequency response M^(/) measured on the NIST

heterodyne measurement system and reported in this calibration includes the power sensor

calibration factor and impedance mismatch, and is given by

In the above expression is the indication ofthe power meter after zeroing and calibration

against the 50 MHz reference signal using a calibration factor of 100%. C is a fimction of the

electrical vector reflection coefficients ofthe power sensor and the photodiode but does not need

to be known. The normalized combined frequency response may be quoted in decibels as 201og[l(/)]. All NIST calibrations report the normalized response in 201og[M(/)] format. The

electrical bandwidth of the Reference Receiver is where 201og[R(/)] falls by 3 dB from the

low frequency level.

If the power sensor and photodetector are permanently attached, the combined normalized

frequency response is fiirther normalized to give 0 dB at 50 MHz. For more information on how

to apply this calibration in a ratio measurement system to sources with arbitrary modulation depth

consult P. D. Hale, C. M. Wang, R. Park, and W. Y. Lau, "A transfer standard for measuring

photoreceiver frequency response," J. Lightwave Technol. 14, 2457 (1996) and NIST $P 250-

xwx.

Procedure

The power sensor was zeroed with a Model XXXX power meter. The power from each laser to the photodiode was adjusted to give 50 mV on the bias monitor, which was assumed to correspond to 50 |iA. Swept scans were made with data points spaced by about one tenth to one

Page: Page x of y

Calibration No.: xxxxx

Date of Report: montii xx, 199x 43

c

m

<idc>RL

(6)
fifth ofthe customer requested resolution. Four scans (two in each direction) were made for each

fi*equency range with overlap ofabout one or more gigahertz to verify consistent operation.

Measurements at specific fi-equencies were extracted fi*om the swept data by a kernel smoothing

technique. For fi-equencies below 1 GHz, data may have been acquired using the heterodyne

system in phase-locked mode; in that case only 3 data sets were taken and averaged directly. The

laboratory temperature and relative humidity were measured at the begining ofeach measurement.

Uncertainty in NIST measurements

Type A uncertainties in the NIST measurement of the fi*equency response of a receiver and

attached power sensor include measurement repeatability, offset correction uncertainty for

transfer standards not referenced to the 50 MHz calibrator, and system environmental effects. The type B uncertainties in the NIST frequency response measurement include scaling and offeet

uncertainties in the rf power meter, drift in the optical power, bias current measurement, and

background noise in the power sensor. A summary of the uncertainty calculation for the

measurement is given in Table 1 . The rf power meter used for these calibrations is a Model

XXXX. For the uncertainties associated with the power meter, the manufacturer's root sum of

squares (rss) specifications were used and assumed to be la in most cases. Since the heterodyne

system uses a swept measurement, different scans will not repeat exactly the same fi*equency. The

response and its standard error at the requested fi-equencies is found using a kernel smoothing

technique. Other uncertainties were estimated empirically. For fiarther details on the uncertainty

analysis for this calibration, see NIST SP 250-x\xx,

For the Director, Report prepared by:

National Institute of Standards

and Technology

Thomas R. Scott, Group Leader Paul D. Hale

Sources and Detectors Group Physicist

Optoelectronics Division Sources and Detectors Group

Page: Page x of y

Calibration No.: xxxxx

Date of Report: month xx, 199x 44
Table 1. Summary of typical frequency response measurement uncertainties for photodiode

with rf power sensor and 3 dB attenuator.

Source of uncertainty Uncertainty,

k=\, %

Type A

Measurement repeatability s <0.05

Offset correction Sg 0.007

Environment effects s^ifm GHz) 0.106 + 0.00064/+

0.00024

Type B

Meter range scaling 0.5

Meter offset 0.023

Optical power drift (power matching) 0.06

Bias current measurement 0.08

Power sensor noise, @ 0.05 GHz 0.025

@-3 dB point 0.05

Total uncertainty (rss), @ 0.05 GHz 0.52

@ -3 dB point, 31 GHz, 4 scans 0.63

Expanded uncertainty (coverage factor k = 2),@ 0.05 GHz 1.05 (0.05 dB)

@ -3 dB point, 31 GHz, 4 scans 1.25 (0.05 dB)

Page: Page x ofy

Calibration No.: xxxxx

Date of Report: month xx,199x 45
gp '(^3) AiuiB^JGOUfi
Appendix F. Shipping instructions

Equipment should be shipped in well-padded foam, or otherwise mechanical-shock insulated

cases, appropriate for reshipment back to the customer. Operation instructions or instruction

manuals should be included, as well as the customer chosen set-up parameters for instrument

functions, including bias voltage, bias current, frequency points required, which connections

(if any) should be left unchanged, and any other special instructions. The customer should

include all cables and connectors that are necessary to calibrate the equipment as specified.

Depending on the power meter model and manufacturer, it may be necessary to include the

power meter and instruction manual.

47

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« I -3 ^ ° S £

a>

_ j= S ' ,0

Q

g ^ 2 fe 13 ^ . § oa 2 o ^ ^ Z c« ro pa o a.