CALIBRATION OF IMPULSE NOISE GENERATORS

Gerome R. Reeve

Electromagnetics Division

Institute for Basic Standards

National Bureau of Standards

Boulder, Colorado 80302

October 1973

Final Report

U.S. Army Contract 67-95896

Prepared for

U.S. Army Electronics Command

AMSEL-NL-C

Fort Monmouth, New Jersey
I
NBSIR 73-343

CALIBRATION OF IMPULSE NOISE GENERATORS

Gerome R. Reeve

Electromagnetics Division

Institute for Basic Standards

National Bureau of Standards

Boulder, Colorado 80302

October 1973

Final Report

U.S. Army Contract 67-95896

Prepared for

U.S. Army Electronics Command

AMSEL-NL-C

Fort Monmouth, New Jersey

U.S. DEPARTMENT OF COMMERCE, Frederick B. Dent. Secretary

NATIONAL BUREAU OF STANDARDS. Richard W Roberts Director

CONTENTS

Page

I. HISTORY OF PROJECT 1

A. Phase I. Dec. 1966 - June 1968 1

B. Phase II. Jan. 19 70 - June 19 7 2 1

II. TECHNICAL REPORT 2

A. Definitions 2

B. Possible Methods of Calibration 3

1. Standard Impulse Generator Method ^

2. Measured Baseband Impulse Method 5

3. Measured Spectral Intensity Method 5

C. Accomplishments of Phase I 8

D. Accomplishments of Phase II 9

1. Continuous Frequency Coverage 9

2. 30 MHz and 1 GHz Radiometers 13

3. 1 GHz Thermal Noise Sources 21

4. Miscellaneous Work 24

E. Observations on Impulse Generators 25

1. Mechanical Switches 25

2. Mercury Wetted Switches 26

3. Solid State Generators 26

III. PROPOSED COURSE FOR FUTURE WORK 28

A. Planning Considerations 28

B. A Newly Proposed System for Calibrating Impulse

Generators 31

iii
CONTENTS (Continued)

Page

C. Proposed Standard Impulse Spectra Generators-- 33

D.. Summary 33

IV. ACKNOWLEDGMENTS = = 39

V. REFERENCES _______ 40

LIST OF FIGURES

Figure 1. SDC Radiometer Block Diagram, 30 MHz.-- - 41

Figure 2. SDC Radiometer Block Diagram, 1 GHz. - 42

Figure 3. Typical Swept Response with Small Frequency

Increments (Bandwidth Determination, Receiver

. B). - 43

Figure 4. Bandwidth vs. Frequency, 0.015-2.0 MHz,

Receiver B. 44

Figure 5. Bandwidth vs. Frequency, 1.5-100 MHz,

Receiver B. 45

Figure 6. Bandwidth vs. Frequency, 1.5-1000 MHz,

Receiver B. 46

Figure 7. Gain vs. Frequency, 0.015-2.0 MHz, Receiver B. 47

Figure 8. Gain vs. Frequency, 1.5-100 MHz, Receiver B.-- 48

Figure 9. Gain vs. Frequency, 1.5-1000 MHz, Receiver B.- 49

Figure 10. Null Sensitivity of Radiometer at 30 MHz,

1 dB change at 40 dB above 1 yV/MHz , 1 kHz

Repetition Rate, (10 minor vertical div. =

1 volt) . -- 50

Figure 11. Null Sensitivity of Radiometer at 1 GHz, 1 dB

change at 40 dB above 1 yV/MHz , 1 kHz Repetition Rate, (10 minor vertical div. = 1 volt).- 51

Figure 12. Null Sensitivity of Radiometer at 30 MHz, 1 dB

change at 30 dB above 1 yV/MHz , 100 Hz Repetition Rate, (10 minor vertical div, - 0.2 volts) 52

iv

j

i
LIST OF FIGURES (Continued)

Page

Figure 13. Null Sensitivity of Radiometer at 1 GHz, 1 dB

change at 40 dB above 1 yV/MHz , 100 Hz Repetition Rate, (10 minor vertical div. = 0.2 volts) 53

Figure 14. Theoretical Useful Operating Range of

Radiometer. 54

Figure 15. Useful Range of Operation, Measured on Breadboard Radiometer at 30 MHz. 55

Figure 16. Sum and Difference Correlation Radiometer

30 MHz and 1 GHz Phase II. 56

Figure 17. 1 GHz Hot and Cold Thermal Noise Sources 57

Figure 18. Interior of 1 GHz Hot Thermal Noise Source 58

Figure 19. Interior of 1 GHz Cold Thermal Noise Source 59

Figure 20. Impulse Produced by Avalanche Transistor Comb

Generator. 60

Figure 21. Fourier Transform of Impulse of Figure 20. 61

Figure 22. Schematic of Avalanche Transistor Impulse

Generator. - - 62

Figure 23. Output of Developmental Solid State Impulse

Generator, Wright Patterson AFB . 63

Figure 24. Proposed Automated Impulse Calibration System

-- Block Diagram. 64

Figure 25. Plot of % Phase Corrections vs. t, 3-5 rf

Cycles. 65

Figure 26. Plot of % Phase Corrections vs. t, 14-16 rf

Cycles. 66

LIST OF TABLES

Table I. Difference in Impulse Response at Band Overlap Frequencies, Receiver A.-- 67

Table II. Performance of 1 GHz Waveguide Beyond Cutoff

Attenuator.-- - - 68

Table III. Percent Phase Correction for Synchronous

Operation. 69

Table IV. Parameters of Calibration Signals 71

V
CALIBRATION OF IMPULSE NOISE GENERATORS

This report covers work performed by the

Electromagnetics Division, Institute for Basic

Standards, National Bureau of Standards, on a Signal Corps contract to develop a calibration

system for impulse noise generators. Various

technical approaches are discussed with their

respective problems. A prototype system which

was constructed is presented along with some

measurements of its capability. Finally another

approach is suggested utilizing newer techniques

and instrumentation which should come closer to

achieving the desired level of performance.

Key words: Impulse; impulse noise; impulse noise

generator, impulse generator calibration.

vi
I . HISTORY OF PROJECT

A . Phase I. Dec. 1966 - June 1968

In September 1966 the U.S. Army Electronics Command, Ft.

Monmouth, N.J. (USAECOM) issued a technical requirement (SCL9035) for a calibration standard for impulse noise generators.

The Engineering Division of the Radio Standards Laboratory

of the National Bureau of Standards responded with a proposal

which resulted in the issuing of a contract at the end of

December 1966 for services to "... consist of developing and

setting up a National Facility for Calibration of the Spectral

Intensity of Impulse Generators in accordance with Technical

Requirement SCL-9035 . . . ." This contract was funded in the

amount of $100,000 and was to run through June of 1968.

With the expiration of the allotted time and funds in

June 1968, work on the project halted and a report (NBS

Report 9743 -- Standards and Measurement of Impulse Spectral

Intensity) was issued covering the accomplishments and outlining what work yet needed to be done.

B . Phase II. Jan. 1970 - June 197 2

In January 19 70, the USAECOM continued the funding of the

original project in an amount of $108,400 and extended the time

I
period to run through September 1971. By February 1971, the

funds were expended while the technical objectives were not

reached. In July 1971 the funding was supplemented with

$25,000 from USAECOM and $25,000 from the Electromagnetics

Division of the National Bureau of Standards. Because of

other commitments work on this effort did not resume until

October 19 71 and continued until June 1972. Although the

objective of creating a "calibration service" was not reached,

progress was made toward this goal as described in the following section.

II. TECHNICAL REPORT

A. Definitions

The problem of defining the quantity of impulse spectral

intensity has been discussed in the previous reports covering

phase I of this project [1,2]. The following is only a brief

summary and the reader is referred to the references cited for

a fuller discussion.

Mathematically, the impulse spectral intensity, V(f), has

been considered to be proportional to the absolute value of

the Fourier transform of one of the following:

(1) a single pulse

;

(2) a finite train of identical pulses;

(3) an infinite train of identical pulses.

2
Number CI) is usually chosen as the representation since

the results of (2) depend on the length of the train of pulses

and (3) has some mathematical difficulties in obtaining a rigorous solution. A more serious problem for engineers is

the constant of proportionality mentioned above. Depending

upon the circuit model chosen and the mathematical technique

used it is possible to justify 1, 2, or /2". This has caused

no end of misunderstanding and confusion in the marketplace.

Since the resolution of these questions must involve a common

understanding and a consensus acceptable to the engineering

community, it more logically falls within the purview of the

cognizant committee of professional engineering societies.

B . Possible Methods of Calibration

There are several methods which can and have been used

to calibrate impulse generators each having particular advantages and disadvantages. These are listed below; while the

list is not exhaustive, it should help to put the work of

this project into better perspective.

1. Standard Impulse Generator Method.

(a) Standard Base Band Impulse Generator. Theoretically it is possible to build a device to generate

an impulsive type of signal* whose characteristics

*In this report the word impulse or impulse signal is used to

refer to the type of signal which is produced by commercial

impulse generators. Typically it consists of a train of narrow (<1 nanosecond) pulses with a repetition period between

10 and 10 ^ seconds.

3
are completely calculable from the physical constants of the generator. Such a generator could

consist of a section of ideal transmission line

charged to a known dc voltage, and an ideal switch.

From the line and switch parameters, and the voltage,

the shape of impulse can be calculated. By means

of the Fourier transform, the spectral intensity as

a function of frequency may be calculated. Using

this known generator as a standard, unknown generators

could be compared to it using an acceptable comparator receiver. In practice, however, it is extremely

difficult if not impossible to build a generator

whose output can be accurately calculated from the

physical constants of the generator. At higher

frequencies, > 0.1 GHz, parasitic reactances and

losses cause perturbations in the output spectrum

which become progressively worse as the frequency

increases

.

(b) Standard Pulsed RF Generator. Instead of constructing a base band impulse generator it is possible

to pulse modulate a CW rf signal to create a flat

impulse spectrum over a bandwidth wide enough to

cover a receiver bandpass centered at or near the

original CW rf signal frequency. The characteristics

of the pulsing signal, and the strictly rf parameters

of the switch can be determined by steady state

4
measurements. While the output pulse is not computed from physical parameters, it is related to

other more easily made electrical measurements of

the generating device. Again the spectrum can be

calculated using the Fourier transform and an unknown

generator compared using an arbitrary receiver. This

method has been implemented in commercially available

equipment although mismatch errors are not accounted

for [3]

.

2. Measured Baseband Impulse Method.

In this approach the unknown impulse is measured by

sampling techniques, e.g., a sampling oscilloscope, the

data points digitized, and using the Fourier transform

the spectral output is computed. Although this method

is not new, recent advances in digital computers and

sampling techniques have made it more practical. It

was one of the methods used by the Power Section, Electromagnetics Division, NBS, to calibrate a solid state

standard impulse generator [4] . Further refinements

of the sampling and digitizing methods were made by

the Pulse and Time Domain Section for a system to measure

passive components by time domain techniques [5].

3. Measured Spectral Intensity Method.

(a) Voltage Measurement. This method makes use

of the impulse response properties of certain types

5
of networks where the peak response of the network

to a CW signal can be related to the peak response

to an impulse signal, by means of the impulse

bandwidth of the network. A receiver whose impulse

bandwidth has been measured is calibrated with a known CW signal whose frequency is set to the peak

response of the receiver. The amplitude of the CW

signal is adjusted to give the same peak response

from the receiver detector as an unknown impulse.

This amplitude combined with the receiver impulse

bandwidth is taken to be a measure of the magnitude

of the impulse signal at the frequency to which the

receiver is tuned. To calibrate an impulse generator as a function of frequency, this procedure must

be repeated at many points over the frequency range

One variation of this method is to use a fixed,

stable, high Q network, followed by a broadband

receiver (even an oscilloscope) as the detector.

This permits the network impulse response to be

more closely controlled although separate networks

would be required for each frequency to be measured

This general method is probably the most used

procedure for calibrating impulse generators because it is relatively easily made and does not

necessarily require sophisticated or hard to obtain

6
equipment. It is the only method, other than the

fourier transform of the base band impulse, using

readily available equipment, that will give the

spectral intensity as a function of frequency over

a large frequency range in a reasonable amount of

time. However, there are several shortcomings to

it, both practical and theoretical, as pointed out

by M.G. Arthur [6]

.

(b) Power Measurement. Instead of measuring the

spectral intensity directly in volts/Hz, it is possible to measure the spectral density in watts/Hz

by radiometric means. The unknown impulse is compared

against a thermal white noise source whose spectral

intensity can be calculated very accurately. Then

by knowing the repetition rate and the internal

source impedance of the impulse generator it is

possible to calculate its spectral intensity [7].

It is not necessary to know the bandwidth of the

radiometer. This method is believed to be potentially

the most accurate since modern radiometers can make

very precise measurements which are then related

to the physical properties of the thermal noise

sources used as standards. Because of the complexity

of the equipment, it is basically a narrowband technique, the measurement being limited to the design

constraints of the radiometer.

7
NBS chose to concentrate on methods 3(a) and 3(b) using

the former to obtain the relative value of the spectral intensity as a function of frequency and the latter to establish

the absolute value of this function at two selected frequencies,

C. Accomplishments of Phase I The accomplishments of phase I have been discussed earlier

[1] but are repeated here in outline form.

1, A sum and difference correlation (SDC) impulse

radiometer was breadboarded for 30 MHz and partially breadboarded for 1 GHz, Test data taken

at 30 MHz indicated good resolution although the

dynamic range was limited. Only preliminary data

were taken at 1 GHz. Block diagrams of these

systems appear in figures 1 and 2.

2. A programmable, swept EMC receiver covering 15 kHz1 GHz (hereafter referred to as receiver A*) was

obtained for use as a relative spectral intensity

measuring device which would be absolutely calibrated at 30 MHz and 1 GHz by the radiometric

method. The unit purchased was one of the first

production models and although preliminary testing

indicated the unit was suitable for the purpose

*A11 commercial equipment referred to in this report is coded

in compliance with NBS policy. For further information contact the Office of Director, National Bureau of Standards.

8
intended, it was plagued with minor mechanical and

electrical faults. In fact the unit was only fully

operational during the last three months of this

contract period.

3. Some of the theoretical aspects of defining impulse

spectral intensity were examined. Although not all

of the questions were resolved, the final report

indicated which direction seemed to offer the most

promise to put the definition on a firmer mathematical basis.

D . Accomplishments of Phase II

1. Continuous Frequency Coverage.

(a) Early in phase II, during further testing of

receiver A, it was discovered that its bandwidth

and its gain were a function of the frequency

to which the receiver was tuned. For a receiver

to be used in measuring spectral intensity, it is

desirable that either its gain or bandwidth not be

a function of frequency. It is relatively easy to

calibrate out variations in one of these parameters

(particularly gain) if the other remains constant.

The manufacturer's specifications for receiver A

implied a constant bandwidth but such was not the

case

9
Since an impulse generator with known output

versus frequency was not available, the receiver was

measured at frequency values that were common to

adjacent bands. For instance, a measurement at

25 MHz on the high end of one band was compared to

a measurement at 25 MHz at the low end of the next

higher band. The gains at the two frequencies were

first adjusted to be the same by measuring a constant

CW signal. Then, if the bandwidths were the same,

two measurements of the same impulse signal should

have been the same, any difference being proportional

to the bandwidth difference. This experiment disclosed variations of up to 4.8 dB at band edge frequencies due to changes in impulse bandwidth (see

table 1) . At lower frequencies this is probably

due to mistracking of overly high "Q" preselector

stages while at very high frequencies differences

in input VSWR between the two bands may also be

a factor. Calibrating the bandwidth as a function

of frequency was considered unprofitable. Besides

being a tremendous amount of work, there was no

guarantee that it would stay constant as the tracking

might change with time.

10
(b) This led to a search for a better receiver to

satisfy the continuous frequency coverage requirement

of the project. Since frequency tracking of the input stages was one of the suspected causes of the band

width changes, attention was focused on broadband

input receivers. As mentioned earlier, the Power,

Current and Voltage Standards Section of the Electromagnetics Division, NBS, had recently completed a project for the Army/Navy/Air Force JTCG-METCAL/

Calibration Coordinating Group which involved the

construction of a solid state standard impulse

generator. As part of the testing of the generator,

they evaluated and used a newly developed, commercial,

spectrum analyzer as an impulse receiver. This

unit, hereafter referred to as receiver B, seemed to

offer enough promise of constant bandwidth and improve

input VSWR that a thorough evaluation was warranted.

The bandwidth of receiver B was measured at

selected points over the frequency range of 15 kHz

to 1 GHz. A frequency synthesizer was used as a source while the receiver was allowed to sweep,

tracing out its own selectivity curve on an X-Y

plotter. The synthesizer was then stepped in frequency about 1/5 the bandwidth and another sweep

taken. This process was repeated many times giving

11
rise to a series of curves as in figure 3. Since

the frequency steps were accurately known, these

furnished a calibration which could then be applied

to any one of the traces and, using numerical integration techniques, its equivalent impulse bandwidth

computed. A receiver bandwidth setting of 3 kHz was

used from 15 kHz to 2 MHz and 300 kHz from 1.5 MHz to

1 GHz. The results of these tests appear in figures

4, 5, and 6. The bandwidth variation was less than

± 2 1/4% over the entire range. The estimated

measurement error was ± 1.51.

The gain was also measured at selected points

over the range. As expected the receiver was not

flat but the gain variation was less than ± 8%, much

less than has been experienced with typical EMC

receivers. The gain plots are presented in figures

7, 8, and 9.

(c) The one penalty that must be paid for these

excellent characteristics in receiver B is that

there is an upper limit to the amplitude of the

base band impulse that can be accommodated without

compression and this limit is necessarily lower

in a receiver with no input tuning. For a 0 . 35

nanosecond wide pulse this upper limit was measured at 70 dB above 1 yv/MHz. This limits

the dynamic range of the instrument to about

12
20 dB for usable signal to noise ratios although with

digital averaging techniques this can be increased

by 20 dB or more. A more complete description of a proposed system using this device is contained in

a later section of this report.

Unfortunately receiver B on which these tests

were conducted did not belong to the project and was

not available to be incorporated into a permanent

calibration setup. Funding for the acquisition of

another receiver had not been planned or budgeted,

therefore it was decided to limit the effort to

completing the work remaining on the sum and difference correlation radiometer at 30 and 1,000 MHz.

Without a receiver of the required quality, there

was no means for obtaining the relative spectral

intensity over the desired frequency range.

2. 30 MHz and 1 GHz Radiometers.

(a) At the end of phase I, a breadboard model of

a 30 MHz radiometer had been constructed and some

work had been started toward a 1 GHz unit. During

phase II the system was completely rebuilt for

increased stability and sensitivity. The sensitivity to a 1 dB change in impulse intensity for two

different repetition rates is shown in figures 10

,

11, 12, and 13. This represents a significant

13
improvement in null sensitivity for a given Integra

tion time over that reported at the end of phase I

[8]. However, the dynamic range of the system was

not increased and it is believed that some of the

changes which were made to improve other parameters

may have decreased the dynamic range slightly,

(b) It is instructive to calculate the useful

operating range of this type of radiometer given

certain assumptions. The assumptions are:

(1) The lower useful limit is where the

impulse power is equal to the thermal noise

generated by the internal impedance of the

impulse generator at room temperature (300 K) .

(2) The total power required to operate the

multiplier (see fig. 1) is 10 mW.

(3) The maximum available output voltage at

the multiplier driver is 10 volts RMS before

compression.

(4) The useful range of the AGC is 26 dB . The

multiplier elements (thermocouples, thermistors

or bolometers) cannot be matched closely enough

to insure adequate linearity. For this reason

the multiplier must be operated at a constant

total power input from each channel which is

14
maintained by an AGC loop. Other requirements

of the radiometer demand phase matching within

± 1.5° in the two channels. Although the AGC

elements constructed have a 35 dB range with

< 1.5° phase shift tracking error, some of

this range must be allowed as a safety factor for

gain drift, aging, etc. This reduces the useful

operating range of the AGC to something less than

30 dB and 26 dB was chosen as a conservative figure.

The calculation is given as follows : For the

minimum useful input signal we have

[V rf)]^

kTB = —^ B (1)

2Rt

where k is Boltzman's constant,

T is the absolute temperature (300 K)

,

B is the bandwidth,

V^(f) is the spectral intensity,

R is the real part of source impedance of impulse

generator, 50 ohms in this example,

T is the repetition period of the impulse signal,

/2 RTk —p— where F is the repetition frequency of the impulse signal. Substituting values,

-9

r T 4.^ /o 0.643 x 10 V (f) m volts/Hz =

s /F

15
0 .643

= (20 log^Q 643 - 10 log^Q F) dB above 1 pV/j^^

= 56.16 - 10 log^Q F

= 26.16 dB above 1 V^/y^^ ^^'^ ^ 1 ^Hz repetition

rate .

If 10 mW is required at the multiplier, 2.5 mW

will be supplied by the impulse signal at the mini

mum sensitivity point. (The unknown signal which

in this case is 1/2 white noise and 1/2 impulse

signal is nulled against an equal CW reference

power. Also in this simplified analysis the effective input noise temperature of the radiometer

which is less than 290 K is ignored.) Therefore

G (max. required gain of radiometer) =

^.j.^ —

G = 0.201 X 10-^^ or 133.05 dB

or G* = 4.483 x 10^ voltage gain assuming a

50 9. input and output

impedance

.

If the maximum signal at the output is 10 volts

then the maximum input signal without AGC is:

^ = 2.231 X 10"^ volts.

Since the bandwidth of the radiometer is 30 kHz

this corresponds to:

16
2.231 X 10"^ X — J- = 7.44 x lO'^ V/MHz

3 X 10^

= 74.4 yV/MHz

= 37.4 dB above 1 yV/MHz.

This is the maximum input that can be handled

before AGC. It also corresponds to the minimum

level at about a 80 Hz repetition rate.

As AGC is applied the maximum impulse signal

that can be handled is increased rapidly until the

limit imposed by the AGC operating range is reached.

Since the AGC must be set so that it begins to act

at the lowest useful signal, the maximum useful

signal can never exceed the minimum useful signal

by more than the AGC range.

However other factors further limit the useful

operating region. There is an upper limit determined

by the bandwidth of the radiometer. Since the bandwidth must be greater than the repetition rate, we

have arbitrarily set the upper limit of the repetition rate at 5 kHz for a 30 kHz bandwidth.

Another constraint on the AGC time constant also

restricts the lower repetition rates which can be

handled. Since the 10 volt output chosen in this

example represents 2 watts in a 50 ohm load, this

could overload the operating point of the multiplier

by 2000/10 or 200 times should a transient system

17
oscillation occur. The only practical way to

protect the multiplier elements from damage is to

make the AGC loop fast enough to reduce the gain

of the system to a safe point. On the other hand

it cannot have too fast a time constant or at slow

repetition rates it will fail to integrate the output

of the AGC power detector and hold the average

power input to the multiplier constant. Thus a

compromise is necessary between available output

swing, AGC time constant, and minimum impulse

repetition rate. A rate of 50 Hz which will accommodate the majority of mechanical impulse

generators was selected.

Figure 14 contains a plot of the various factors

which limit the range of the SDC radiometer and its

useful operating region. By comparing this with

figure 15 which is a plot of the region where the

sensitivity of the breadboard radiometer appeared

optimum at 30 MHz, reasonable agreement between the

predicted and actual operating region is observed.

The experimental data were taken below 50 Hz by

lengthening the AGC time constant although this

is not considered a safe mode of operation. As

mentioned earlier the rebuilt radiometer constructed

18
during phase II did not appear to have as great a dynamic range as predicted although time did not

permit detailed measurements of this parameter,

(c) 1 GHz Radiometer.

(1) The 1 GHz radiometer constructed during

phase II did not operate nearly so well as the

30 MHz unit. Besides reduced sensitivity caused

by the higher effective noise temperature of the

radiometer (which was expected) there was an

instability problem possibly due to relative

phase changes in the two channels. The effect

of phase instability on the null condition of

the radiometer is a function of the cosine of

the phase difference between the signals in

the two channels. A change in relative phase

will cause an apparent change in the value of

the signal being measured. It is believed that

some of the shielding enclosures used to contain

leakage from the various oscillators and

amplifiers acted as high "Q" cavities at 1 GHz

and that room temperature changes acting on

these enclosures produced significant phase

shifts in various internal leakage signals.

Due to the termination of the project for lack

of funds these problems could not be investigated

and solved.

19
(2) In addition to the above-mentioned problem

and the usual troubles in suppressing leakage,

it proved extreme ly difficult to construct an

adequate waveguide beyond cutoff attenuator for

the 1 GHz reference GV signal. Input and output

matching and mechanical stability problems were

major obstacles and the attenuator became a small

research project in itself. Although they were

ultimately solved and a satisfactory attenuator

constructed, the data for which appear in table

II, the effort required was much greater than had

been planned and reduced the time available for

the rest of the project.

A photograph of the combined 30 MHz and

1 GHz radiometers appears in figure 16.

(d) Measurements with the SDC Radiometers. Although,

at the cessation of work in June 1972, the radiometers

were not fully evaluated some comparisons were made

with the NBS standard impulse generator. A comparison of the readings at two different levels at a

1 kHz repetition rate is given below:

20
(Levels given in dB above 1 yV/MHz.)

Standard Impulse Computed from

Generator Setting SDC Radiometer

30 MHz

1 GHz

36 ---37.15

31 33.69

34 35. 35

29 31.00

The estimated error of the standard impulse generator

is ± 1/2 dB. Since no error analysis was made for

the radiometer and the values shown represent only

a single determination, no firm conclusions can be

drawn. It is interesting to note that the radiometric

measurement is higher by 1 to 2 dB and a possible

reason for this will be given in a following section

in which solid state impulse generators are discussed.

3. 1 GHz Thermal Noise Sources.

In order to use the SDC radiometer to calibrate

an unknown generator it is necessary to have standard

thermal noise generators at two different temperatures.

The internal impedance of these generators must be adjustable to match the impedance of the unknown generator.

Two thermal noise sources were already available at

30 MHz, but it was necessary to construct two more thermal

noise sources at 1 GHz. The two temperatures were chosen

21
to match those of the 30 MHz units. They were boiling

water or 373 K and the boiling point of liquid nitrogen

(LN2) , or 77 K. The method chosen to tune the generators

internal impedance is believed to be novel for this type

of noise source. Varactor diodes were placed at predetermined locations along the transmission line from

the source resistor to the outside connector but close

enough to the resistor that they would be immersed in

the temperature stabilizing medium. The itself acted

as the cooling medium in the cold source. In the hot

source a high boiling point (431 K) , fluorochemical

fluid* with a high thermal capacity was used. The

temperature was maintained by a servo system using a thermistor sensor and heating resistors. Since all the

possible noise generating lossy elements, including

the basic source resistor, the tuning varactors , and

most of the transmission line, are held at a known

constant temperature the noise output can be accurately

predicted. The internal temperature of both sources was

monitored with precision platinum resistance thermometers

Figure 17 is an external view of the two sources

while figures 18 and 19 show some of the internal construction of the hot and cold source respectively.

*Manufactured by Company "C".

22
The dials on the top of the sources control the

voltages applied to the varactors for tuning. At 1 GHz

the cold source can match any impedance within a 1.2/1

VSWR circle around the center of a 50 Z Smith Chart

o while the hot source can match any impedance within a

1.3/1 VSWR circle.

The systematic uncertainties were evaluated for

the two sources and are presented in the table below.

The uncertainties are expressed in terms of temperature

(Kelvins)

.

Source of Uncertainty 77 K Source 575 K Source

1. Platinum resistance 0.066 K 0.11 K

thermometer calibration.

2. Thermal conduction along 0.005 K 0.005 K

thermometer readout leads.

5. Calibration of thermometer 0.0001 K 0.001 K

readout bridge.

4. Temperature gradients 0.01 K 0.01 K

in baths

.

5. Transmission line 0.15 K 0 . 10 K

correction uncertainty.

Total uncertainty 0.2511 K 0.226 K

Corrections for the electrical loss in the part of

the transmission lines not immersed in the bath were

calculated using a linear thermal gradient model. The

23
corrections are listed below while the uncertainties in

the corrections have been included in line 5 above.

77 K Source 575 K Source

Correction due to electrical +0.262 K -0.105 K

losses in transmission line.

4. Miscellaneous Work.

In testing the system during its development it

was necessary to have a stable, repeatable, impulse

source of variable repetition rate. While the better

mechanical types may be repeatable, their repetition rate

is not variable. A developmental solid state impulse

generator was obtained on loan from Wright-Patterson

Air Force Base and was used for all of phase I and part

of phase II. During phase II it had to be returned, and

since the NBS standard impulse generator referred to before

was not available on a day to day basis, it was necessary

to find a substitute. An inexpensive, solid state comb

generator for frequency calibration work had been developed

by another group at NBS [9]. Since it was basically an

impulse generator with about a 500 picosecond width

whose repetition rate could be varied, a Fourier transform analysis was performed on its output. The impulse

and the Fourier transform are shown in figures 20 and 21.

As can be seen the output is relatively constant dropping

24
only 11 dB at 1 GHz. While this is not as flat as the

better mechanical generators it is more than adequate for

many purposes. The absolute output was not measured

since it was not of great importance but was estimated

to be 70 dB above 1 yV/MHz. A schematic for the generator

appears in figure 22. The significance of this data is

that they demonstrate that a solid state impulse generator

usable to 1 GHz is not difficult or expensive to build.

The cost of parts for the impulse generator (not including

the PRF driving source or power supplies) is under $50.

Any good laboratory pulse generator, or square wave

generator with adequate rise time will drive it.

E . Observations on Impulse Generators

During the course of developing the measurement system

many different types of impulse generators were employed as sources. Although the information which follows is mainly

qualitative in nature it may be helpful to those unfamiliar

with the instruments that are currently available.

1. Mechanical Switches.

Generators in this category generate impulses by

discharging a section of transmission line with an

electromechanical switch. The switch is driven by a

low frequency oscillator (50-2000 Hz) which is sometimes

25
adjustable. Impulses produced by such an instrument are

subject to both phase and amplitude jitter and perhaps

long term drift due to mechanical wear. In general they

are not useful for semi -precision measurements nor worthy

of precision calibration.

2. Mercury Wetted Switches.

These generators are similar to those described above

except that the switch contacts are mercury wetted and

the repetition rate is fixed and limited to 60 Hz or less.

The absence of contact bounce and long term wear reduces

the amplitude jitter and drift to an acceptable level

although phase jitter is still present. This type of

generator is the one most commonly in use today, either

as a separate calibrator or built into EMC receivers as

an internal calibrator. Although small mechanical variations will cause significant variations in output from

unit to unit, they are individually worthy of precision

calibration. Their phase jitter may make certain calibration schemes, e.g. those using sampling techniques,

less useful.

3. Solid State Generators.

These generators use charge storage or step recovery

diodes to generate a fast voltage transition. Such

diodes retain stored carriers for a very short time after

26
the terminal voltage changes from forward to reverse bias

.

When the charge is finally depleted an extremely rapid

(as short as 50 ps) transition takes place from the

short circuit to the open circuit condition. The rapid

voltage rise across the diode is shaped by passive elements

into a short pulse. Impulse generators using these devices

produce spectra which have negligible amplitude jitter

while phase jitter depends primarily upon the stability

of the driving PRF generator. Accurate synchronizing

signals are therefore available. A photograph of an

oscilloscope display of the output of the solid state

generator from Wright Patterson AFB referred to earlier

is shown in figure 23. Either positive or negative

output is available.

There is a problem which sometimes arises with

solid state impulse generators . Because the impulse

generating device must be driven at the required repetition rate by other circuitry, the driving signal waveform

can leak through the step recovery diode circuit and

appear attenuated at the output of the generator.

Some of this leakage signal may be very close in time to

the desired impulse in which case it appears as additional ringing on the output waveform. But other

signals may appear much later in time, particularly if

the driving waveform is a square wave. The amplitude

27
o£ these extraneous signals may be quite small, < 1%

of the amplitude of the main impulse, so that they are not

often observed on an oscilloscope. If they occur after

the response time of an EMC receiver, they will not produce

a change in its peak reading meter display. But since

their time duration may be quite long, their energy content

approaches that of the main impulse at lower frequencies

.

Since a radiometer responds to the total energy within

its pass band, not to the largest amplitude of the signal,

the radiometric measurement would always indicate a

higher value than measurement techniques which deal only

with the main impulse and/or a receiver's response to it.

This might be the cause of the measurements reported

earlier with the SDC radiometer being about 1 dB greater

than expected.

III. PROPOSED COURSE FOR FUTURE WORK

A. Planning Considerations

Several conclusions can be drawn from the work so far

accomplished. First, it would appear that radiometric techniques, while having potentially high accuracy, are not

desirable for general measurements of impulses. Radiometers

measure total power and respond to artifacts between the major

impulses which means they will be measuring something different

28
than the typical impulse receiver which responds only to the

peak amplitude o£ the largest signal appearing at its detector.

It seems reasonable that a calibration system should measure

that parameter of the impulsive signal which is most commonly

useful for making measurements by other equipment. Further,

when the impulsive noise source is at very low repetition

rates, the power contained in the pulse train compared to the

available power from the generator's internal source impedance

results in poor signal power to noise power ratios. A

radiometric technique would be of great value in cross-checking

the accuracy of another type of impulse calibration system

providing the impulse signal source used as a transfer standard is known to be free of artifacts and the repetition rate

is kept high to maintain an adequate signal to noise ratio.

Second, if an amplitude response system is to be used,

it should be employed only as a means for comparing impulse

generators since the theory that relates the receiver's absolute response to an impulse is based on measuring a quantity

called impulse bandwidth which is not related rigorously to

other more easily measured quantities. To develop a calibration service from this technique an impulse signal source

must be developed that can qualify as a reference standard,

which means its output must be calculable from physical

constants and/or other electrical measurements. A baseband

rectangular pulse generated by an electronic switch could be

29
a candidate at low frequencies where its time duration can be

accurately measured by digital techniques and its amplitude

measured with the switch in the "on" position using D.C. instruments . At higher frequencies the RF pulse technique appears

to be the only candidate which can satisfy the same criteria;

namely, be calculable from its steady state electrical measurements. The criteria for selecting the operating ranges

and characteristics of these generators are discussed in

section C below.

Third, the measured value of the spectral intensity of

available impulse generators is generally a function of frequency and depends in part on the reflection coefficients of

the generator and calibration system. Therefore, measurements

of these quantities must be made at closely spaced intervals

in the frequency domain and appropriate corrections made.

This necessitates that any practical calibration system must

be automated, using computer control in the measurement process.

Actually the mismatch problem affects the usage of the

impulse generator in every application. It can be handled

as mentioned above during the calibration, but it is almost

universally ignored when the generator is used to calibrate

other equipment such as field intensity receivers. A receiver which has been thus "calibrated" with an impulse

generator is then connected to an antenna with unknown

30
mismatch errors further degrading the usefulness of the original

calibration. These problems concerning the application of

impulse generators are beyond the scope of this report and

are only mentioned to show that the problem is far from trivial.

B . A Newly Proposed System for Calibrating Impulse Generators

The former plan of using a SDC radiometer and thermal

noise standards as a basis for providing a calibration service

for impulse noise sources needs to be discarded and replaced

with a more applicable and viable system. This section presents such a plan. The plan is an outgrowth of the work

performed and the lessons learned from this effort.

Figure 24 shows a generalized block diagram of the newly

proposed impulse calibration system. The basic components are

a programmable receiver with low input VSWR and constant bandwidth (such as receiver B evaluated during this contract) , a tracking generator which provides a CW signal which is automatically tuned to the center of the receiver's passband, a precision switch with on-off ratios > 70 dB , a pulse generator,

an impulse synchronized peak sample and hold detector with

digitized output, an automatic network analyzer (to measure the

complex reflection coefficients of the generators and the

receiver), suitable RF switches, and a mini -computer.

31
The system would function in the following manner. The

automatic network analyzer would first measure the complex

reflection coefficients of the standard generator, the unknown

impulse generator, and the measurement receiver at as many

frequencies as desired under the direction of the mini -computer

This data would then be stored in the computer memory. Next

the standard generator would be applied to the input of the -

receiver. This will be a baseband pulse for frequencies up to

1 MHz and a pulsed RF source for frequencies above 1 MHz. When

the pulsed RF source is used, the RF source for the switch

comes from the tracking generator, thus assuring the spectrum

is centered with respect to the receiver. The computer tunes

the receiver over the frequency range making measurements at

the frequencies where the reflection coefficients were measured.

The peak sample and hold device samples the output of the

detector at the proper time to obtain the peak of the detected

envelope. The instant of sampling is determined by a trigger

pulse from the broadband directional coupler and a delay

circuit which accounts for the receiver response time. The

output of the sample and hold circuit is digitized and stored

in the computer. To improve signal to noise ratios, from 100

to 10,000 measurements can be taken at each frequency and the

computer performs a signal averaging, storing the averaged value

before moving to the next frequency. When the measurements

are finished, the computer calculates the measured spectral

32
intensity relative to the standard, makes the necessary corrections to the raw values based on the reflection coefficients,

and prints out the corrected values of spectral intensity vs.

frequency. In this brief description many minor steps necessary

to the operation of the automatic network analyzer have been

left out since they are incidental to the measurement process.

C . Proposed Standard Impulse Spectra Generators

A rectangular baseband pulse of the type to be discussed

produces a spectrum given by the Fourier transform [10]

X(f) = At sine fx (2)

, . sin irfT where sine = ^ .

A

Since we will be dealing with pulse repetition rates slow

enough that the measurement system response has essentially

died away before the next pulse appears, it is legitimate to

use the transform of a single pulse rather than the Fourier

series. The sine function is = 1 for small values of the

argument and is reduced by 1% when the argument is = 0.24

and is reduced by 5% for an argument = 0.55. Hence for very

33
short pulses (t very small) , the impulse spectral intensity

S(f) is approximately equal to Ax. This expression,

S(f) = At , is commonly taken as the definition o£ impulse

spectral intensity sometimes with the addition of a coefficient which we will assume is 1. The difficulty with impulse

generators using this principle is that in an effort to keep

the spectrum flat to as high a frequency as possible t is made

so short that neither the shape nor amplitude can be predetermined and are extremely difficult to measure accurately.

If it is desired that a spectrum be flat to 5% to 1 MHz, the

situation improves greatly. The pulse width necessary is

.55 = fxir, (3)

= 'ih ' b = 175 nanoseconds.

This is a pulse length which can be generated and measured

with an accuracy of better than II. Using the fastest switching

logic, rise and fall times can be less than 1.5 ns and the

amplitude under the steady state "on" condition can be measured very accurately. Corrections for the 5% rolloff at 1 MHz

can be computed along with second order corrections for the

finite rise and fall times. Also since one pulse response

must die away before the next pulse appears, there is an upper

limit to the pulse rate. For the shape of passband in receiver

B it is necessary for the PRF < ^^i^^widthresponse

to have decayed to 1% of the peak value [11]. As a rule of

< BW thumb it is safe to make the PRF = ^—^

34
Above 1 MHz the switched or pulsed RF signal is more

advantageous. The spectrum for this signal is [12]

X(£) = 1/2 At sinc(£-£^)T

± 1/2 At sinc(£+f^)T (4)

£or a rectangular shaped pulse, where £^ is the carrier frequency. Basically the spectrum shape is the same as that o£

a baseband pulse translated to the carrier frequency and symmetrical about it. The ± before the second part of the expression depends on whether a cosine or sine function is chosen

to represent the carrier signal.

Since we are really interested in frequencies about the

center o£ the spectrum, f^, the expression can be rewritten

X(f') = 1/2 At sinc(± f'T)

± 1/2 At sinc(2f^ ± f')T (5)

where f is the frequency difference measured from the carrier

Looking at eq. (5) and comparing it with eq. (2) shows

that the first sine function will contribute most of the

expression's value at frequencies near the carrier while the

second sine function, sometimes called the "phase correction"

35
[13] , will become significant at frequencies well removed from

the carrier. The requirement for one percent spectrum rolloff

on either side of the carrier can be found from the first sine

function, e.g., for 150 kHz on either side of the carrier

(the widest bandwidth of receiver B) a pulse < 0.51 y second

is required. The pulse repetition rate requirement is the

same as for the proposed baseband pulse generator.

However, it is possible to choose the value of x so that

a whole number of rf cycles are included in the pulse (synchronous operation) or a nonintegral , irrational number of cycles

(asynchronous operation) . Since this will affect the spectrum

through the magnitude of the phase correction, it is necessary

to examine these conditions in more detail. Figure 25 is a plot of the phase correction expressed as a percentage of the

basic "sine" function, i.e.

sinc(2f ±f')T

% Phase correction = 100 x . (6)

sinc(f'T)

The carrier frequency is 30 MHz and x was chosen to cover the

region between 3 and 5 integral cycles of the carrier. The

curve represents the percent correction at a separation f

from the carrier of -150 kHz which is 1/2 the widest bandwidth

of receiver B. The correction for f = +150 kHz is of comparable magnitude. Figure 26 is the same type of plot except

X has been chosen to cover between 14 and 16 rf cycles. From

these two plots it is evident that the spectrum amplitude can

36
vary in asynchronous operation as a function of the number of

rf cycles in the pulse with the variation decreasing as the

number of rf cycles in the pulse increases. If at least 15

cycles are included in the rf pulse the variation will be less

than 1.1% at ± 150 kHz, the widest bandwidth of receiver B.

This means that at 30 MHz and above asynchronous operation

is feasible if the pulse width is =0.5 y second. From 1 to .

30 MHz the pulse could be lengthened. This would also require

a narrower bandwidth since the spectrum would fall off faster

on each side of the carrier. Synchronous operation offers

some advantages however. Table III shows the percent phase

correction for integral numbers of cycles at various carrier

frequencies and at different spacing, f , from the carrier.

This table shows that at a given f^, the phase correction is

independent of the number of cycles in the rf pulse providing

these are exactly a whole number of cycles. Keeping the criterion

of a phase correction less than 1.11 and spectrum rolloff

within the receiver bandwidth, the following chart can be

made showing the parameters of the calibration signals.

37
D. Summary

The calibration system which has been outlined represents

what is believed to be a practical, state-of-the-art approach

to the problem. It differs from the system originally proposed

in that a radiometric measurement is not made on the unknown

generator. While this represents some change in philosophy, it

would not be possible except for the tremendous progress which

has been made in automated measurements and in stable, programmable laboratory receivers and spectrum analyzers . Despite the

fact that the project did not result in a completed set of hardware for the system described, enough has been learned to point

the way to a practical calibration approach capable of being

used at frequencies up to 1 GHz and in principle even in the

microwave region.

38
IV. ACKNOWLEDGMENTS

The author wishes to express his thanks to the following

persons who at one time or another were connected with the

project

:

M.G. Arthur, for developing the background and theory

of the SDC radiometric measurements while project

leader during phase I.

L.D. Driver, for acting as project engineer during extensive periods of phase II and for developing the 1 GHz

thermal noise standards

.

E.E. Baldwin, for constructing much of the electronic

equipment used during the project.

A.R. Ondrejka and L.D. Martin, for making the measurements on the receivers which were evaluated.

E.B. Larsen, for offering many useful suggestions on RF

pulse calibration.

39
V. REFERENCES

[I] M.G, Arthur, "Standards and measurement o£ impulse

spectral intensity," Unpublished NBS Report, Oct. 1969.

[2] M.G. Arthur, "Impulse spectral intensity calibration

at the National Bureau of Standards, Proc. 1969 Dept.

Defense Electromagnetic Compatability Symposium, pp.

133-138.

[3] "Calibration of impulse spectral intensity using pulsed

CW signals," Andy Hish Associates, Tech. Report 0-057010-

1, 5 pages, 19 71.

[4] A.R, Ondrejka, "Development and evaluation of a solid

state impulse generator," Unpublished NBS Report.

[5] W.L. Cans and N.S. Nahman, "Fast fourier transform implementation for the calculation of network frequency domain

transfer functions from time domain waveforms," NBS

Interagency Report 73-303, Final Report DoD Calibration

Coordination Group Contract 72-65.

[6] Op. cit. [1] pp. 4-5.

[7] Op. cit. [1] pp. 24-35.

[8] Op. cit. [1] , p. 45.

[9] J.R. Andrews, "A frequency calibrator for UHF using an

avalanche transistor," QST , Vol. LVI , No. 5, pp. 16-18,

May 19 72 .

[10] A.B. Carlson, "Communication systems, an introduction to

signals and noise in electrical communication," McGraw

Hill, p. 31, 1968.

[II] "Spectrum analysis -- pulsed RF," Application Note 150-2

Hewlett Packard, p. 11, Nov. 1971.

[12] Op. cit. [10] , p. 39

.

[13] "The frequency spectrum for a tone burst," Instrument

Notes IN-105, General Radio Company, July 1965.

40
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Figure 16. Sum and Difference Correlation Radiometer,

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56

58

Figure 20. Impulse Produced by Avalanche Transistor

Comb Generator.

60

INPUT

O+250V

IMPULSE

OUTPUT

>-aT<

CI - 47 pF, dipped mica.

C2 - 1500 pF, 500 V feedthrough.

DLI - 50-ohm microstrip transmission line.

DL2 - 50-ohm microstrip transmission line.

Ql - 2N3904

Rl, R2 - 100 ohms, 1/8 W.

R3 - Two 100-ohm, 1/8 W resistors in parallel

R4 - 100,000 ohms, 1/2 W.

RFCl - Ferrite bead on R4 lead.

Ul - TTL inverter, 7404.

Figure 22. Schematic of Avalanche Transistor Impulse Generator.

62
1 ns

Positive Pulse

2V

i

T

1 ns Negative Pulse

Figure 23. Output of Developmental Solid State Impulse

Generator, Wright Patterson AFB .

63
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f
66
Table I. Difference in Impulse Response at Band Overlap

Frequencies, Receiver A.

Bands Overlap Frequency

1, 2 30 kHz

2 , 3 60 "

3, 4 120 "

4, 5 250 "

5 , 6 500 "

6 , 7 1.1 MHz

7, 8 2.4 "

8, 9 5 MHz

9 , 10 11 "

10 , 11 25 "

11, 12 50 "

12, 13 100 "

13, 14 200 "

14, 15 500 "

Difference (Abs . Value)

0.7 dB

2.8 "

0 " 1.8 " 1.0 " 0.9 " 2.3"

3.0 " 1.2 " 3.7 " 1.5 " 2.0 " 4.8 " 4.5 "

67
Table II. Performance o£ 1 GHz Waveguide Beyond Cutoff

Attenuator.

Attenuator Dial Actual Departure From

Indication Attenuation Linearity

dB dB dB

0 0 0

1 0.993 +0.016

2 1.983 +0.029

3 2.969 +0.038

4 3.950 +0.042

5 4.936 +0.051

For dial readings greater than 5 dB, the actual attenuation

change is determined by multiplying the indicated change by

the factor 0.977.

A ^ T = 0 .977 A. ,. , actual indicated

All measurements were made by setting the dials in the direction of increasing attenuation. The repeatability of the

attenuator is estimated to be ± 0.001 dB and the backlash of

the dials and gearing is less than ± 0.005 dB

.

The insertion loss of the attenuator assembly (0 dB position) is 4 3.4 dB. This includes the 6 dB pad on the output.

The overall accuracy of calibration is ± 0.05 dB out to a setting of 30 dB.

68
Table III. Percent Phase Correction for

Synchronous Operation.

£ = 30 MHz

c

15 cycles

10 cycles

5 cycles

10 cycles

5 cycles

2 cycles

f

t % Correction £

t 1 Correction

+ 150 kHz 0 .250 -150 kHz -0 .250

50 kHz 0 .083 - 50 kHz -0 .083

15 kHz 0 .025 - 15 kHz -0 .025

5 kHz 0 .008 5 kHz -0 .008

1 . 5 kHz 0 .003 -1.5 kHz -0 .002

+ 150 kHz 0 . 2 50 - 150 kHz -0 .250

50 kHz 0 .083 - 50 kHz -0 .0 83

15 kHz 0 .025 - 15 kHz -0 .025

5 kHz 0 .008 - 5 kHz -0 .008

1 . 5 kHz 0 .00 3 -1.5 kHz -0 .002

+ 150 kHz 0 .250 - 150 kHz -0 .250

50 kHz 0 .083 - 50 kHz -0 .08 3

15 kHz 0 .025 - 15 kHz -0 .025

5 kHz 0 .008 - 5 kHz -0 .008

1 . 5 kHz 0 .00 3 -1.5 kHz -0 .002

£

c

= 10 MHz

+ 150 kHz 0 .745 -150 kHz -0 . 755

50 kHz 0 .250 - 50 kHz -0 .250

15 kHz 0 .075 - 15 kHz -0 .075

5 kHz 0 .025 - 5 kHz -0 .025

1 . 5 kHz 0 .008 -1.5 kHz -0 .007

+ 150 kHz 0 .745 -150 kHz -0 . 755

50 kHz 0 .250 - 50 kHz -0 .250

15 kHz 0 .075 - 15 kHz -0 .075

5 kHz 0 .025 - 5 kHz -0 .025

1.5 kHz 0 .008 -1.5 kHz -0 .007

+ 150 kHz + 0 . 745 -150 kHz -0 .755

50 kHz + 0 .249 - 50 kHz -0 .250

15 kHz + 0 .0 75 - 15 kHz -0 .0 75

5 kHz 0 .025 - 5 kHz -0 .025

1.5 kHz 0 .008 -1.5 kHz -0 .007

69
Table III. Continued

f = 3 MHz

c

10 cycles

5 cycles

2 cycles

10 cycles

5 cycles

2 cycles

£

t ^ Correction £

t % Correction

+ 150 kHz 2 .439 -150 kHz -2 .564

50 J\.l 1 0 82 7 - 50 kH7 - 0 8 40

15 kHz 0 .250 - 15 kHz -0 .250

5 kHz 0 .083 - 5 kHz -0 .083

1.5 kHz 0.025 -1.5 kHz -0.025

+ 150 kHz 2.439 -150 kHz -2 .546

50 kHz 0 .827 - 50 kHz -0 .840

15 kHz 0 .025 - 15 kHz -0 .250

5 kHz 0 .083 - 5 kHz -0 .083

1 . 5 kHz 0.025 -1.5 kHz -0 025

+ 150 kHz . 2 .439 -150 kHz -2 .564

50 kHz 0.827 - 50 kHz -0 .840

15 kHz 0 .250 - 15 kHz -0 .250

5 kHz 0 .083 - 5 kHz -0 .083

1 5 kH7 n 0 25 - 1 5 kHz -0 02 5

£ = 1 MHz

+ 150 kHz 6.977 -150 kHz -8.108

'^O 7 4 3Q - 50 kH7 - 2 564

15 kHz 0 .745 - 15 kHz -0.755

5 kHz 0 .250 - 5 kHz -0 .250

1.5 kHz 0 .075 -1.5 kHz -0 .075

+ 150 kHz 6 .977 -150 kHz -8.109

50 kHz 2 . 4 39 - 50 kHz -2 .564

15 kHz 0 . 745 - - 15 kHz -0 .755

5 kHz 0 .250 - 5 kHz -0 .250

1.5 kHz 0 .075 -1.5 kHz -0 .075

+ 150 kHz 6 .977 -150 kHz -8.108

50 kHz 2 .439 - 50 kHz -2 .564

15 kHz 0 .745 - 15 kHz -0.755

5 kHz 0 .250 - 5 kHz -0 .250

1.5 kHz 0 .075 -1.5 kHz -0 .075

'^0

I

l'
Table IV. Parameters of Calibration Signals

f

c Pulse Width Mode Receiver BW Max. PRF

> 30 MHz 0.5 y sec. asynch. 300 kHz 150 kHz

10-30 MHz 0.1666-0.5 y sec. synch. 300 kHz 150 kHz

(5 cycles)

3-10 MHz 0.5-1.666 y sec. synch. 150 kHz 50 kHz

(5 cycles)

1-3 MHz 1.666-5 y sec. synch. 30 kHz 15 kHz

(5 cycles)

< 1 MHz 0.175* y sec. N.A. 300 kHz 150 kHz

(baseband pulse)

*5% rolloff at 1 MHz for which correction is made.

71
FORM NBS-n4A (1-71) U.S. DEPT. OF COMM. 1. PUBLICATION OR REPORT NO. 2. Gov't Accession

BIBLIOGRAPHIC DATA No. SHEET NBSIR 73-343

3. Recipient's Accession No.

4. TITLE AND SUBTITLE

Calibration of Impulse Noise Generators

5. Publication Date October 1973

6. Performing Organization Code

7. AUTHOR(S)

Gerome R. Reeve

i. Performing Organization

9. PERFORMING ORGANIZATION NAME AND ADDRESS

NATIONAL BUREAU OF STANDARDS, Boulder Labs.

Boulder, CO 80302

10. Project/Task/Work Unit No. 2726266

11. Contract/Grant No. U.S. Army Contract

67-95896

12. Sponsoring Organization Name and Address U.S. Army Electronics Command

AMSEL-NL-C

Fort Monmouth , New Jersey

13. Type of Report & Period Covered „.Final Report

Dec. 1966 - Jimp. 1972

14. Sponsoring Agency Code

15. SUPPLEMENTARY NOTES

16. ABSTRACT (A 200-word or less factual summary of most significant information. If document includes a significant

bibliography or literature survey, mention it here.) This report covers work performed by the Electromagnetics Division , Institute

for Basic Standards, National Bureau of Standards, on a Signal Corps contract to develop a calibration system for impulse noise generators . Various technical approaches

are discussed with their respective problems . A prototype system which was constructed

is presented along with some measurements of its capability . Finally another approach is

suggested utilizing newer techniques and instrumentation which should come closer to achieving the desired level of performance

17, KEY WORDS (Alphabetical order, separated by semicolons)

Impulse; impulse generator calibration; impulse noise; impulse noise generator.

18. AVAILABILITY STATEMENT

UNLIMITED.

r~, FOR OFFICIAL DISTRIBUTION. DO NOT RELEASE

TO NTIS.

19. SECURITY CLASS

(THIS REPORT)

UNCLASSIFIED

20. SECURITY CLASS

(THIS PAGE)

UNCLASSIFIED

21. NO. OF PAGES

22. Price

USCOMM-DC 66a44-P7l

USCOMM - ERL