NiST

PUBLICATIONS

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NIST SPECIAL PUBLICATION 400-86

U.S. DEPARTMENT OF COMMERCE/National Institute of Standards and Technology

aoo-86

1990

C.2
NATIONAL INSTITUTE OF STANDARDS &

TECHNOLOGY

Researdi Infcmnatiofi Center

Gakhersburg, MD 20899

DATE DUE

Demco. Inc. 38-293
Semiconductor Measurement Technology:

Thermal Resistance Measurements

Frank F. Oettinger and David L. Blackburn

Semiconductor Electronics Division

Center for Electronics and Electrical Engineering

National Engineering Laboratory

National Institute of Standards and Technology

Gaithersburg, MD 20899

July 1990

U.S. DEPARTMENT OF COMMERCE, Robert A. Mosbacher, Secretary

NATIONAL INSTITUTE OF STANDARDS AND TECHNOLOGY, John W. Lyons, Director
National Institute of Standards and Technology Special Publication 400-86

Natl. Inst. Stand. Technol. Spec. Publ. 400-86, 78 pages (July 1990)

CODEN: NSPUE2

U.S. GOVERNMENT PRINTING OFFICE

WASHINGTON: 1990

For sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, DC 20402-9325
Semiconductor Measurement Technology:

THERMAL RESISTANCE MEASUREMENTS

Table of Contents

Page

I. POWER TRANSISTORS 1 Introduction 1 The Concept of Thermal Resistance 2

The Effect of Nonuniform Junction Temperature on the Definition of Rqjr ... 2

The Variation of Rqjr with Device Operating Conditions (/c?^CJs) 3

General Methods for Measuring Device Temperature 7

Control of the Thermal Environment 13

Selection of Temperature-Sensitive Electrical Parameters 14

Measurement of Temperature- Sensitive Electrical Parameters 17

Safe-Operating-Area Limits 21

DC Thermal Limit 21

Second Breakdown Limit 26

Relating the Chip Temperature to the Measured Temperature 29

Determination of Area of Power Generation and Peak Junction Temperature . . 29

Detection of Onset of Current Constrictions 33

Reasons for Measuring Temperature 33

Measurement of Integrated Power Devices 38

Power Transistor Thermal Standardization Activities 41

II. INTEGRATED CIRCUITS 41

Introduction 41

Design of Thermal Test Chips 43

Computer Simulations 46

Thermal Properties of Materials 46

Guidelines for Thermal Test Chip Design 46

Utilization of Thermal Test Chips 56

Mounting Considerations 56

Direct Measurement Procedures 60

Indirect Measurement Procedures 60

IC Thermal Standardization Activities 66

Acknowledgments 68

References 68

iii
List of Figures

Page

1. Idealized versus real- world temperature distribution for a power transistor ... 4

2. A 20-W power transistor coated with a thermographic phosphor shown for

three different operating conditions, each dissipating 16 W 6

3. Measured thermal resistance as a function of collector current at a constant

power of 20 W for a 35-W triple-diffused transistor 8

4. Temperature distribution of a Darlington power transistor measured with an

automated infrared microradiometer 9

5. Photomicrograph of a Darlington transistor in the previous figure after

experiencing a high power pulse 10

6. Heating and measurement waveforms as they appear at device terminals . . . .12

7. Schematic of the temperature control system and temperature-controlled heat

sink used at the National Institute of Standards and Technology 15

8. Schematic of measurement circuit for using standard method for measuring

temperature of a bipolar transistor 16

9. Examples of calibration curves for each of the power MOSFET TSEPs 18

10. Schematics of measurement circuits for each TSEP 19

11. Measured and corrected emitter-base voltage waveform during measurement

of the temperature for a bipolar transistor .22

12. Measured T versus y/i and extrapolated temperature for a bipolar transistor . . 24

13. A representative example of a manufacturer's published safe-operating-area

(SOA) limits for a power transistor 25

14. The manufacturer-specified SOA limits and the actual SOA limits determined by

an infrared microradiometer for a power transistor which has no specified

second breakdown limit .27

15. The manufacturer-specified SOA limits and the actual SOA limits determined by

an infrared microradiometer for a device that is inadequately rated in the

second breakdown region by the manufacturer 28

16. The temperature distribution of a power MOSFET measured with an automated

infrared microradiometer 30

17. The electrically measured cooling response of a power transistor versus the

square root of time for two different operating conditions for the same power . . 32

18. The manufacturer-specified SOA limits and points along the measured SOA

limits as determined by an infrared microradiometer, by the peak temperature

electrical technique, and by the standard electrical technique 34

19. The measured Veb at 20 /xs after the removal of a 1-s power pulse versus Vce

for a number of values of /c 35

20. Schematic of Darlington transistor 39

IV
21. Pictorial representation of heating and sensing areas on thermal test chips ... 44

22. Photomicrographs of representative thermal test chips with inserts showing

sensing diode circuit configuration 45

23. Three-layer structure modeled by computer code 47

24. Influence of temperature on the thermal conductivity of integrated-circuitrelated materials 48

25. Influence of chip size on the junction-to-case thermal resistance of a three-layer

package structure 51

26. Influence of heat source layout on chip surface temperature for test chip arrays

and single cell chips 52

27. Influence of chip size on internal power dissipation for a junction-to-case

temperature difference of 20 °C 53

28. Influence of heat source layout on chip surface temperature for a silicon chip

eutectically bonded to an alumina substrate 54

29. Comparison of computer-simulated and measured thermal resistance for a

leadless ceramic chip carrier package 55

30. Package mounting arrangements 58

31. Thermal response measurements for an alumina chip carrier package with an

integral aluminum heat sink for various mounting and cooling configurations . . 61

32. Example of a temperature-controlled heat sink 62

33. Example of an infrared microradiometer 63

List of Tables

Page

1. Comparison of Thermal Resistance Measurement Techniques for a Bipolar

Transistor 5

2. Comparison of Thermal Resistance Measurements for a MOSFET 20

3. Procedure for Detecting Nonthermal Switching Transients and

Extrapolating Temperature to Instant of Switching 23

4. Advantages of Thermal Response Testing for Die Attachment 37

5. Procedure for Measuring the Temperature of Darlington Transistors 40

6. Power Transistor Thermal Test Method Standardization 42

7. Thermal Conductivity of Materials 49

8. Fabrication Technologies for Thermal Chips 57

9. Computer Simulations of the Thermal Characteristics of Various

Chip- Package Configurations 59

10. Precision and Accuracy of Infrared Measurements 64

11. IC Thermal Test Method and Specification Standardization 67

V

Semiconductor Measurement Technology:

THERMAL RESISTANCE MEASUREMENTS

Frank F. Oettinger and David L. Blackburn

Semiconductor Electronics Division

National Institute of Standards and Technology

Gaithersburg, MD 20899

Abstract

This Special Publication reviews the thermal properties of power transistors and integrated circuits and discusses methods for characterizing these properties. The discrete

devices discussed include bipolar transistors and metal-oxide-semiconductor field-effecttransistors. Measurement problems common to these devices, such as deciding the reason

for requiring a particular measurement, adequate reference temperature control, selection

of a temperature-sensitive electrical parameter, and separation of electrical and thermal

effects during measurement are addressed. Due to the inherent difficulties in measuring and analyzing the thermal properties of active integrated circuits, an approach using

specifically designed thermal test chips for evaluation of new die-attachment and packaging

schemes is finding wide acceptance in the industry. In this Special Publication, indirect

(i.e., electrical) measurements, direct (e.g., infrared) measurements, and computer simulation techniques for thermally characterizing integrated circuits are discussed in terms of

their usefulness in characterizing VLSI packages.

Key words: integrated circuits; junction temperature; power transistors; safe-operating- area limits; temperature-sensitive electrical parameter; thermal measurements; thermal

resistance; thermal test chips.

Disclaimer: Certain commercial equipment, instruments, or materials are identified in this

paper in order to specify adequately the experimental procedure. Such identification does

not imply recommendation or endorsement by the National Institute of Standards and

Technology, nor does it imply that the materials or equipment identified are necessarily

the best available for the purpose.

I. POWER TRANSISTORS [1,2]

Introduction

Accurate characterization of the thermal properties of power transistors is critical to the

reliability of the systems using these devices. Failure of a single power device can shut

down a computer, bring to a halt a motor-driven system, or stop a vehicle dead in its tracks. Power devices are often expected to run hotter than other components, but the

excessive temperature rise of an inherently "bad" device that "slipped through" screening

will lead to early, often catastrophic failure. Likewise, the absence of adequate thermal

characterization information may cause an otherwise "good" device to be used in a circuit

1
that stresses it beyond its thermal limits.

The purpose of this section is to review the thermal properties of power transistors and to

discuss methods for characterizing these properties. The devices discussed include bipolar

transistors and metal-oxide-semiconductor field-effect-transistors (MOSFETs). Measurement problems common to these devices, such as deciding the reason for requiring a particular measurement, adequate reference temperature control, selection of a temperaturesensitive electrical parameter (TSEP), and separation of electrical and thermal effects during measurement are addressed. Finally, the needs for thermal characterization of evolving

devices such as high voltage and power integrated circuits and merged bipolar/MOSFET

devices are mentioned.

The Concept of Thermal Resistance

The concept of thermal resistance has been developed over the years as an aid to device

manufacturers and users for calculating the junction temperature of operating devices.

The concept is based upon an analogy between the electrical and thermal properties of

materials, with temperature, heat flow, and thermal resistance being analogous to voltage,

current, and electrical resistance, respectively. Thus, the equation

RejR = ^ p

^

(1)

where

R-ejR — thermal resistance between the junction and the reference point (°C/W),

Tj^Tr = temperatures of the junction and the reference point, respectively, (°C), and

P — power dissipated (W),

are used to define the device thermal resistance.

In applying the electrical analogy to the thermal problem and in using eq (1), it is implicitly

assumed that

1) the junction temperature is spatially uniform, and

2) RejR is independent of the device operating conditions, i.e., independent of the

values of collector current Ic and collector-emitter voltage Vce used to obtain the

power P[P ^ Ic X Vce)-

Neither of these assumptions is valid for actual power transistors though, and care is needed

in applying the results of eq (1) to real devices. The effect of these invalid assumptions on

the defined and measured Rqjr is discussed in the remainder of this section.

The Effect of Nonuniform Junction Temperature on the Definition of Rqjr - The idealized

isothermal temperature distribution upon which eq (1) is based is shown superposed upon a

2
more realistic nonisothermal temperature distribution in figure 1. The calculation of Rqjr

using eq (l) is straightforward for the isothermal case, but it is not immediately obvious

how to apply the equation, or even if it is applicable, for the nonisothermal real- world case. This is because Tj is not well defined for the latter case in that it takes on a continuum

of values between the peak junction temperature and the minimum junction temperature.

For the concept of thermal resistance to be applicable to the real-world situation, some

unique and meaningful value for Tj must be defined.

Because it is the peak junction temperature Tj^^p^ak) that is most important for predicting

the reliability and safe operation of a device, Tj^^p^ak) is the most logical choice for use in

place of Tj in eq (1) [3]. Thus, if eq (1) is rewritten as

_ Tj{peak) - Tr

^9JR{peak) = p

where RejR{peak)i peak-junction-to-reference-point thermal resistance (°C/W), a unique

and meaningful value of thermal resistance for any junction temperature distribution is defined. Unfortunately, all of the commonly used electrical methods for measuring junction

temperature indicate some average temperature which is less than Tj^p^ak) by an unknown

amount. Thus, if RejR^peak) is to be specified, some arbitrary correction factor usua^'

must be applied to the measured value of Rqjr to account for the difference between tiie measured junction temperature and the peak junction temperature. There appears to be

no consistency in the application of the arbitrary correction factor in common practice,

and the factor is often insufficient to ensure safe device operation.

The peak junction temperature of power transistors can be estimated using a modification

of the standard electrical measurement method in conjunction with some simple mathematical procedures [4]. Thus, realistic values for RejR{peak) can be determined without the

need for arbitrary correction factors. A comparison of the actual peak thermal resistance

as determined using an infrared microradiometer, that determined using the methods described in reference [4], and the average thermal resistance determined using the standard

electrical technique is shown in table 1 for a number of devices and for a number of operating conditions. As can be seen in the table, a considerable diff'erence may exist between

the measured average thermal resistance and the peak thermal resistance.

Unfortunately, as also can be seen in table 1, the value of RejR{peak) depends upon the

device operating conditions. This is true even for different values of Ic and Vce which

give nearly the same (or even the same) value of power [Ic x Vce)- Thus, even though a unique value of thermal resistance can be defined for a given set of operating conditions,

it is unique only to those specific operating conditions.

The Variation of Rqjr with Device Operating Conditions {Ic-, Vce) ~ The value oi RgjRf^p^ak)

depends upon Ic and Vce because the distribution of /c throughout the device, and therefore the distribution of the power density, depends upon the device operating conditions

[5,6]. An example of this can be seen in figure 2 for a 20-W planar power transistor which

is dissipating 16 W of power for three different operating conditions. The device is coated

3
POSITION

Figure 1. Idealized versus real- world temperature distribution for a power transistor. (A)

idealized isothermal temperature distribution upon which the concept of thermal resistance

is based; (B) more representative of the real-world temperature distribution for a power

transistor.

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Figure 2. A 20- W power transistor coated with a thermographic phosphor shown for three

different operating conditions, each dissipating 16 W. The phosphor when illuminated with

ultraviolet radiation as done here is brightest at the cooler areas and darkest at the hotter

areas, (a) Ic = 0.8 A, Vce = 20 V; (b) Ic = 0.4 A, Vce = 40 V; {c)Ic = 267 mA,

VcE = 60 V.

6
with a thermographic phosphor [7] which appears darkest at the hottest areas of the device

in this figure. For the low-current, high-voltage conditions {Ic = 267 mA, Vce = 60 V),

the power density is much more localized (as evidenced by the very dark "hot-spot") than

for either of the other two operating conditions. This is despite the fact that the same

power is being dissipated for each condition. The peak thermal resistance for the operating condition depicted in figure 2(c) is about 9.7 °C/W and for the condition depicted

in figure 2(a) is about 9.1 °C/W. Although the RejR{peak) in this instance varies by only

a few percent, variations as a function of operating conditions of several hundred percent

are not uncommon for other devices [5].

In general, for low-current, high-voltage operation, the thermal resistance is a strong function of operating conditions. At the other extreme, high current and low voltage, it is not

as strong a function of operating conditions, but it still may depend upon Ic and VceThis can be seen in figure 3, which shows a plot of thermal resistance versus Ic at constant

power for a bipolar power transistor. The reason for the large values of Rqjr at low values

of /c is that the thermal-electrical feedback mechanisms that initiate current crowding and

eventually second breakdown [8] are stronger at small values of Ic than at large values.

Thus, the current tends to be more constricted and the peak temperature higher for these

conditions. It is shown later that the presence of current crowding and the associated large

RejR{peak) values oi Ic shotdd be taken into account when generating SOA limits.

General Methods for Measuring Device Temperature

There are two sets of "generic" methods in general use for measuring the operating temperature of a semiconductor device. One set requires that the semiconductor chip be exposed

for viewing and in some instances that a foreign substance be placed on the surface of the

chip. Examples are infrared microradiometry [9] and the use of liquid crystals [10,11] and

thermographic phosphors [7]. These methods allow one to create a map of the temperature

distribution on the chip surface. The infrared method is of most interest here for mapping

purposes because it can be used to quantitatively map the surface temperature, whereas

the other methods, although extremely valuable, are more qualitative in nature. The other

set of "generic" methods uses a temperature-sensitive electrical parameter (TSEP) of the

device as a thermometer. These methods can be performed on fully packaged devices but

give a single, average temperature value for the chip.

The infrared methods are the most useful methods for accurately determining the distribution of temperature on the device surface. Because the surface temperature is never

uniform, a single temperature value cannot fully characterize the temperature of the device, and large-temperature nonuniformities can be mapped using the infrared techniques.

An example of the value of this is shown in figure 4, in which the temperature distribution for a power Darlington chip, measured using an automated infrared microradiometer,

is displayed. From this display, it is clear that one area of the chip (the hottest area)

is dissipating most of the power. Much of the chip is apparently dissipating little or no

power. Only from such a temperature map can one establish the nonuniformity of the

temperature and the area of the chip that is dissipating most of the power. The impact

that this has on chip reliabihty is demonstrated by the picture in figure 5 of the same

7
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COLLECTOR CURRENT (A)

Figure 3. Measured thermal resistance as a function of collector current at a constant

power of 20 W for a 35-W triple- diffused transistor.

8
Figure 4. Temperature distribution of a Darlington power transistor measured with an

automated infrared microradiometer.

9
Figure 5. Photomicrograph of a DarHngton transistor in previous figure after experiencing

a high power pulse. Dark area leading from bond wire at bottom center of figure is melted

emitter metallization.

10
chip after it experienced a brief high power ptdse. The darkened area is melted aluminum,

leading from the emitter bond wire to the spot on the chip that was hottest during the

temperature scan. The major problem with infrared methods is that they require that the device being studied

be "decapped" and visible to the infrared. In addition, the methods are time-consuming,

they are quantitatively practical only for steady-state measurements, they require extensive

temperature control equipment for controlling the thermal environment of the device being

studied, and they require expensive infrared measuring equipment. For these reasons, such

measurements are often only made to diagnose problems, such as the one demonstrated

by figures 4 and 5.

Temperature measurement methods that use TSEPs are the most often-used methods.

They have the advantage that they can be performed on fully packaged devices, oftentimes may be done quickly and are easy to perform, can be quantitatively accurate for

steady-state or transient conditions, can often be done with relatively inexpensive electrical equipment, and in certain special circumstances may be done with minimal heat sinking

or thermal environment control. The typical "electrical" measurement of temperature is done in two phases, the calibration phase and the measurement phase. In the calibration

phase, the value of the particular TSEP chosen is calibrated versus temperature. For instance, if the emitter-base voltage of a bipolar transistor is selected as the TSEP, the value

of the emitter-base voltage is determined at various chip temperatures. This is accomplished by externally heating the device, such as in an oven or by a temperature-controlled

hot plate or heat sink, and then measuring the emitter-base voltage with a constant cur- rent passing through the device. The constant current must be very small as it is assumed

that the device is at the same temperature as the oven environment or hot plate which

reqtures that no power (or at least minimum power) be dissipated in the device. In the

measurement phase, the device is self-heated by dissipating at a much larger power than

that at which the calibration was done. To measure the temperature, the device must be

switched from the heating condition (high power) to the measurement condition (the same

low current and power at which the calibration was performed) with a heating power duty

factor of approximately unity. An example of the heating and measurement waveforms for

measuring the temperature of a bipolar transistor is shown in figure 6.

In calculating the power transistor junction-to-reference-point thermal resistance and as- suming the previously stated approximations, i.e., 1) the power dissipation during calibration is negligible and 2) the heating power duty factor is approximately unity, the following

equation is used [12]:

RdJR

_Tj — Tr _ Vmh — Vmc

I AVjvfc ~ P{Avg) ~ Ph \^Tmc (3)

Im'

where

RejR — thermal resistance, junction to reference point (°C/W),

11
0

t

VCE

Figure 6. Heating and measurement waveforms as they appear at device terminals. This

is a specific example for a bipolar transistor.

12
Tj = junction temperature (°C),

Tr — reference point temperature (°C),

Im = measuring/calibration current (mA),

P[Avg) = average heating power appHed to transistor causing temperature differ- ence Tj - Tr (W),

Ph — magnitude of heating power applied to transistor (W),

^MH = value of TSEP corresponding to the temperature of the junction heated

by Ph and measured at Im (mV),

Vmc — value of TSEP during calibration at Im and specific value of Tmc (mV),

Tmc = calibration temperature measured at reference point (°C), and

^VmcI ^Tmc = TSEP temperature coefficient measured at Im (mV/°C).

There are some subtle but potentially major difficulties with the switching requirements

of the electrical methods. One is that the device is cooling during the actual measurement

because the heating power has been removed. For an accurate measurement to be made,

this cooling must be considered and taken into account. Second, the device cannot change

electrical states instantaneously. It takes a finite time for the device to switch fully from

the heating state to the measurement state. If a measurement is made before the electrical

switching is completed, large errors can result in the measured temperature. Finally, as

figure 4 demonstrates, the device has some distribution of temperature, but the electrical

methods indicate a single device temperature that is some average of this temperature

distribution. Methods have been developed to overcome some of the limitations of the

electrical measurement procedures. They are discussed in later sections of this pubhcation.

Control of the Thermal Environment

If one is to measure accurately the thermal characteristics of a device, the thermal environment of the device during measurement must be known and controlled. This environment

will typically not be the environment experienced by the device in its application, but

it must be known and controlled for one to relate the measured characteristics to the

application characteristics.

Knowledge and control of the thermal environment usually means knowledge and control

of the temperature at a specified point on the device or its package. To "know" the

temperature at the reference point means that a thermocouple, thermistor, or thermometer

of some other type be attached to or in some way determine the temperature at the point.

It is usually not satisfactory to know only the temperature of the surrounding gas or fluid

because in most instances the reference temperature and the fluid or gas temperature will

be different.

For steady-state, multiple-pulse, or single-pulse heating conditions greater than about

13
300 ms in length, it is usually not sufficient to use a heat sink that is not temperature

controlled, such as merely a large copper block. The problem is that the uncontrolled heat

sink begins to rise in temperature, heated by the device under test, making it impossible

to maintain a controlled reference point temperature. Control and measurement of the

reference point temperature is usually difficult and requires ancillary equipment such as temperature sense and control units and methods of heating and cooling the heat sink. A

schematic of a temperature-controlled heat sink and temperature control system used at the National Institute of Standards and Technology for many years is shown in figure 7. The copper heat sink is continuously cooled by the chilled water and intermittently heated

by the heating resistors as required. The temperature control unit senses the temperature

with a thermocouple and commands the power supply to supply heating current to the

resistors to maintain the preset temperature. With the system shown in the figure, the

reference point temperature can be controlled to within about 0.1 °C.

For some important measurements, a heating pulse of less than 300 ms is desirable. Such a

case is for die-attach evaluation [13-18]. In this instance, one wants to make the die-attach

portion of the thermal resistance to be as large a portion as possible of the measured

thermal resistance. This is accomplished with a short heating pulse of the appropriate

length. More will be said about this later, but for this type of measurement, close control

of the thermal environment is not critical because the thermal resistance between the

device package and the environment does not enter into the measurement.

Selection of Temperature-Sensitive Electrical Parameters

Although a variety of factors enter into the selection of a TSEP for measuring the temperature of a particular device, the two most important factors are the accuracy with which

the TSEP can determine the device temperature and the convenience or ease of use of

the particular parameter. The particular parameter chosen may also depend upon the reason for performing the thermal characterization. If the purpose is to qualify a device

to some particular rating or limit, the most accurate parameter should be used. If the

purpose is to screen devices quickly or as an outgoing or incoming inspection, then an

easier, more convenient parameter may be chosen, provided good correlation between the

actual temperature and the parameter being measured has been established.

It has been well established that the most accurate and the easiest TSEP to use for bipolar

devices is the emitter-base voltage measured at a small, constant current. In addition, it has been found that the preferred technique to use is the "emitter-only switching" method

for which the collector voltage is not switched between the heating conditions and the

measurement conditions [12,19-20]. A schematic of the measurement circuit is shown in

figure 8 and the various waveforms at the terminals of the device under test are those

shown in figure 6. The situation for power MOSFETs is more complicated than for the bipolar transistor. It has been shown that the most accurate TSEP for the MOSFET is the threshold voltage,

i.e., the source-gate voltage measured at a small drain current [21]. Although the procedure

developed for using the source-gate voltage as the TSEP is nearly identical to that used

14
Chilled H2O

Temperature Control

Controller w ' Output

Heating

Resistor Power Supply

Thermocouple

Input

DUT

DUT Power and Measurement

Circuit

DUT ,,TC Wires

Thermocouple

wires

Emitter & base wires

Heating resistors

Figure 7. Schematic of the temperature control system (a) and temperature-controlled

heat sink (b,c) used at the National Institute of Standards and Technology.

15
16
for the emitter-only switching method used for bipolar transistors, and is equally as easy, a different method has been standardized and is therefore most often used [22]. This

second method uses the forward voltage of the intrinsic drain-body diode present in all power MOSFETs as measured using the drain-source voltage as the TSEP. Examples of

a calibration curve for each of these TSEPs are shown in figure 9. Schematics of the

measurement circuits for these two techniques are shown in figure 10 with a comparison

of the extrapolated TSEP temperatures to that of infrared measurements of the peak chip

surface temperature given in table 2. This method has been selected as the standard

because it was in popular use for several years before the threshold voltage method was

developed and also because the calibration of the diode forward voltage is somewhat simpler

than that of the threshold voltage. This latter condition is related to the fact that the

variation in dV^a/dT for a group of "identical" diodes is usually less than the variation in

dVth/dT for MOSFETs. Thus, rather than generating a single calibration curve valid for

all MOSFETs of a single type, as is done for diodes, the threshold voltage method could

require calibration for each device. Fortunately, because power MOSFETs usually have

a comparatively uniform chip temperature distribution, the standard method is usually

satisfactory.

The situation for merged bipolar/MOSFET devices is not well defined. The only commercially relevant device in this category today is the Insulated Gate Bipolar Transistor

(IGBT). One difficulty with this device is that there is no intrinsic diode available as there

is for the power MOSFET. It is possible that the threshold voltage of this device will yield

similarly accurate temperature measurements as for the power MOSFET and be equally

as easy to use. The other possible candidate as a TSEP is the forward voltage drop, or sat- uration voltage, of the device measured at a low forward current. Little reported work has

been done on comparing the various TSEPs for temperature measurements of the IGBT,

although the threshold voltage has been proposed as the TSEP in a military standard.

Measurement of Temperature- Sensitive Electrical Parameters

The method chosen to measure the TSEP is critical to determining the accuracy and

repeatability with which the TSEP indicates the actual device temperature. Because

the measurement procedure nearly always requires the device under test to be switched

from a high-power, heating condition to a low-power, measuring condition, the presence,

control, and identification of "nonthermal" switching transients are problems for all device

types. Nonthermal switching transients are defined as electrical transients that mask the

temperature variations of the TSEP that one is attempting to measure. Often, these

nonthermal effects are inseparable from the desired thermal effects without some "postdata collection" processing. Ignoring nonthermal switching transients probably is second

only to lack of reference temperature control in contributing to temperature measurement

errors.

Typically, the TSEP has a variation with temperature of a few millivolts per degrees

Celsius. Thus, for an accurate and reHable measurement of temperature, a few millivolts error in the TSEP translates into a few degrees Celsius error in temperature. For instance,

the emitter-base voltage at a low emitter current typically has a temperature variation

17
20 30 40 50 60 70 80 90 100

TEMPERATURE (°C)

Figure 9. Examples of calibration curves for each of the power MOSFET TSEPs.

18
19
TABLE 2

Comparison of Thermal Resistance

Measurement Techniques for a MOSFET

Operating

Conditions Temperature Rise Above Case

Id{A)/Vds{V) IRM VsG Vds

0.5/60 33.5 31.0 28.8

1.0/30 33.5 31.0 28.8

1.5/20 33.5 31.0 27.8

0.5/60 38.0 35.0 36.5

1.0/30 38.0 35.0 36.1

1.5/20 38.0 35.0 36.1

1.0/60 80.0 80.3 70.7

1.5/40 80.0 80.3 70.4

0.5/60 37.0 36.8 36.1

1.0/30 37.0 36.8 36.1

1.5/20 37.0 36.8 35.0

1.5/40 73.0 69.0 62.0

1.0/20 34.3 31.4 28.1

20
of about -2 inV/°C. In the usual switching appHcation, if the emitter base voltage has

settled to within a few tens of millivolts of its final value, the device has for all intents and

purposes finished switching. It may take less than a microsecond for the voltage to settle

to within a few tens of a millivolt of its final value, but it typically takes more than 10 fis for the voltage to settle to within a millivolt of the final value. It has even been shown

that the metallic leads of a typical power device package contribute to these nonthermal

switching transients [23]. An example of this is shown in figure 11. The net result of all this is that to measure the temperature accurately using a TSEP, one must wait until the

nonthermal switching transients have subsided. This will nearly always require a delay of

at least 20 fis after switching to the low-level measurement condition from the high-level

heating condition. Usually, the delay is longer, sometimes being 100 fis or longer. A

method has been developed for detecting the presence of nonthermal switching transients

and for extrapolating the temperature measured after a delay time to the temperature of

the device at the moment that the device was switched [24]. The procedure used is based

upon one- dimensional cooling theory and is summarized in table 3. An example of the

measured temperature, with and without consideration of nonthermal switching transients,

and the extrapolated temperature are shown in figure 12.

Safe- Operating-Area Limits

Device manufacturers pubHsh safe-operating-area (SOA) limits for nearly all of their power

transistors. The limits define an area on an Ic — Vce plane within which safe operation is guaranteed and within which the peak junction temperature is guaranteed not to exceed

the specified safe maximum temperature, Tj{^^ax)- These limits are set for both dc and

pulsed operation, but the Hmits of the SOA of concern here are the dc thermal limit and the

dc second breakdown limit denoted by the line segments 1 and 2, respectively, in figure 13.

The discussion is also limited to forward-bias SOA limits. A comparison of the techniques

generally used for determining the pulsed thermal limits has been published [24]

.

DC Thermal Limit - The dc thermal limit of the SOA is in reality determined from the

thermal resistance of the device. Because the thermal limit defines the limit along which

Tj{peak) = Tjimax), oue finds from eq (2)

[Ic X VCE)LIMIT

where {Ic x Vce) limit defines the locus of points along which Tj^^p^ak) = ^J(max)- If RejR{peak) assumed constant, as are Tjf^^ax)

{Ic X Vce)limit = C, (5)

where C is constant (i.e., constant power), or,

ln[Ic]= -ln[VcE] + C'. (6)

21
T

—I— I I I 1 I I

1 1 T"—I1 r

TIME (gs)

Figure 11. Measured and corrected (for nonthermal switching transients) emitter-base

voltage waveform during measurement of the temperature for a bipolar transistor.
TABLE 3

Procedure for Detecting Nonthermal Switching Transients

and Extrapolating Temperature to Instant of Switching

1. Measure temperature in normal manner but as a function of time after switching.

A series of measurements between 10 fxs and 250 /zs after switching is satisfactory.

2. Plot measured temperature, T, versus the square root of time after switching,

3. Nonthermal switching transients have subsided at the time the plot of T versus \/t

becomes linear.

4. Draw a straight line through the linear portion of the T versus \/i plot and extend

the line to < = 0. The intercept with the t = 0 axis is the extrapolated temperature

at the instant of switching.

An example of the results of this procedure is shown in figure 12.

23
160

150

140

O

- 130

LU

cr

LU

QIJLl 110

100

90

80

5 10

1 T" 20 40

TIME (MS)

80 160 320

120)e^_

Extrapolation of

linear region

T=kti/2

o Temperature from

measured Vg^

8 10 12 14 16 18 20

TIME (ps)

Figure 12. Measured T versus \/i and extrapolated temperature for a bipolar transistor.

The time after switching for each measured temperature is noted on the top scale.

24
LJJ OC

DC

O

OCO

o

o

o

10 c

1.0

0.1

I

I I I I iii| I —I I 1 1 1 1 1| 1—I I 1 1 III

1

- Thermal Limit

2 - Second

Breakdown

Limit

« I I I I I M J » I I 1

1

' ' '

I I I m

1 10 100 1000

COLLECTOR-EMinER VOLTAGE (V)

Figure 13. A representative example of a manufacturer's published safe-operating-

(SOA) limits for a power transistor.

25
Thus, a straight-line relation with a slope of -1 exists between /nfJc] and /n[Vc'£;] if RejR{peak) is Constant. This can be seen in figure 13, where line segment 1 has a slope of

-1. The specified thermal limit of the SOA will usually run from the maximum safe value

of collector current, Ic{max)^ some lower value of Ic at which point a limit of slope less

than -1, the second breakdown limit, commences.

As has been shown previously, RejR{peak) cannot be considered a constant for all device

operating conditions. However, in the high-current, low-voltage operating region (near the

Ic(max) limit), the assumption of a constant RejR{peak) is usually reasonable. An indication

of this can be seen in figure 14 where the manufacturer's published SOA (solid line) as well as the SOA determined from the actual peak junction temperature measured with an infrared microradiometer (dashed line) are shown for a device with no specified second

breakdown limit. Within the specified thermal limit, this device can operate quite safely

with Tj^peaJk) significantly below Tj(^^ax)- addition, the thermal limit as determined by

the infrared microradiometer has a slope of approximately -1, which indicates a constant

RejR(peak)- ^or this device then, the safe operating area is conservatively limited by

the specified Tj(^jnax)- 1^ has also been observed that for most devices operating at or near Ic{max) (and thus limited by the -1 slope of the safe operating curve), TjjK^p^ak) ~

TjR{Avg)i where Tjn^^j^^g^ is measured using the Veb emitter-only switching technique.

This is because of the more uniform temperature distribution at high currents.

Second Breakdown Limit - The second breakdown limit of the SOA is usually determined

by actually permitting a number of devices to enter second breakdown. The limit is usually set at values of 7^7 and Vce somewhat less than those for which second breakdown

occurs in the sample to add a margin of safety. Often, however, the thermal resistance

of the device is not measured over this region because it is assumed that the thermal

limit must continue with a slope of -1 and would thus be outside the second breakdown

limit. As has been described, the low-current, high-voltage operating conditions, for which

the published SOA is usually second-breakdown-limited, RejR(peak) is usually a strong

function of operating conditions. This means that the published high-current, low-voltage

thermal limit probably cannot be linearly extrapolated to the low-current, high-voltage

operating region. In fact, the low-current, high-voltage conditions, RejR{peak) can be so large that the junction temperature exceeds the published safe maximum temperature well

within the second-breakdown-limited safe-operating area. Thus, though the device may

be protected from second breakdown in this region, it is not thermally protected.

An indication of the way variations in RejR{peak) can aff'ect the actual SOA limits of a

device that is severely second-breakdown-Hmited can be seen in figure 15. This figure shows

the manufacturer's published SOA (solid line) as well as the experimentally determined

SOA (dashed line) which was determined by using an infrared microradiometer to estimate

the actual peak junction temperature. Within most of the specified thermal limit (specified

slope of -1), this device can operate quite safely with Tj^^p^^k) significantly below Tj^^max) even though the true slope deviates from -1. For the low-current, high-voltage conditions,

though, the actual thermal limit falls well within even the specified second breakdown

limit, and thus this device should be thermally limited over almost all of its operating

26
5.0

< 2.0

LU

1.0

o

DC

o 0.5

o

LU

-J—

1

8 0.2

0.1

T

1

1 I I I I I i~n I I I 1

1

J I

I 10 20

COLLECTOR-EMITTER VOLTAGE (V)

11

50 100

Figure 14. The manufacturer-specified SOA limits (solid line) and the actual SOA limits

determined by an infrared microradiometer (dashed line) for a power transistor which has

no specified second breakdown limit.

27
5.0

10 20 50 100 200 500

COLLECTOR-EMITTER VOLTAGE (V)

Figure 15. The manufacturer-specified SOA limits (solid line) and the actual SOA limits

determined by an infrared microradiometer (dashed line) for a device that is inadequately

rated in the second breakdown region by the manufacturer.

28
range.

There are a number of electrical measurement techniques that can possibly be used to

determine better the SOA limits for power transistors. These techniques can be used by

manufacturers in determining the original specified limits or by users to determine if a

particular device in a particular application is operating with a safe peak temperature

and safe temperature distribution. These techniques are discussed below along with their

possible appHcations to the determination of SOA limits.

Relating the Chip Temperature to the Measured Temperature

The term "chip temperature" is ambiguous. The semiconductor chip does not have a single

temperature, but has a distribution of temperature. An example of a relatively uniformly

distributed surface temperature for a power MOSFET chip is shown in figure 16. An

example of a nonuniform distribution for a power bipolar transistor is shown in figure 4. The surface temperature distributions of these two chips were measured with an infrared

microradiometer. The electrically measured temperature for the uniform temperature

MOSFET was 94 °C, only 4 °C below the actual peak temperature. The TSEP used

for the MOSFET was the source-gate voltage at a low level current. The electrically

measured temperature for the bipolar transistor of figure 4 was 64 °C, 16 °C below the peak

temperature. For the bipolar transistor with a nonuniform temperature distribution, the

electrically measured temperature is a poor representation of the actual peak temperature.

It is important to be able to estimate the peak junction temperature of a power transistor.

It is equally important to be able to determine when the temperature distribution becomes

nonuniform and thus when the standard electrically measured temperature is considerably

below the peak temperature (electrical techniques typically average the temperature). In

addition, the onset of temperature nonuniformity is usually a precursor to second breakdown and certednly enhances the possibility of premature device failure. One possible

method of improving SOA limits would be to set the limit as determined either by the

peak junction temperature or by the onset of temperature nonuniformity, whichever occurs

first [25]. Because the onset of temperature nonuniformity will generally precede second

breakdown, this SOA limit would generally fall within the usual second-breakdown limit.

Determination of Area of Power Generation and Peak Junction Temperature - It is possible

to estimate the actual area of steady-state power dissipation for a power transistor [24] and

thus determine the extent of current crowding in the device, because for at least the first 250 /is of cooling, power transistors cool from steady-state according to one- dimensional

heat flow theory. That is,

T{t)-T{0) = Kt'/\ (7)

where

29
Figure 16. The temperature distribution of a power MOSFET measured with an automated

infrared microradiometer.

30
K =

2P

y/pirkc Ae ' (8)

£ind

m = temperature at time t (°C),

m = temperature at instant of power removal (°C),

t = time after removal of power (s),

p = power dissipation (W),

p = density (g/mm^),

k = thermal conductivity (W/mm-°C),

c = thermal capacity (W-s/g-°C), and

Ae = area of power generation or effective active area (mm-^)

A plot of temperature of the device as it is cooling versus t^^^ results in a straight line,

and the area of power generation Ae can be found from the slope of that line K. This is demonstrated in figure 17, where the cooling rate of a power transistor is shown for two

different operating conditions, but each for the same power level. The slope of the curve Ic = 0.2 A gives an area of 0.72 mm"^ and the slope of the curve for Ic= 0.5 A gives

Ae = 1.96 mm^. The total chip area is about 6.46 mm"^. It is obvious that the junction

temperature for the 0.2-A condition should be much hotter (same power, smaller area) than

for the 0.5-A condition, and both the infrared and electrical measurements indicate this.

It is important to note, though, that for the small-area case, the actual peak temperature

(as indicated by an infrared microradiometer) is considerably higher than the electrical

indication. The difference is not nearly so great for the large-area case, because, as has

been mentioned, the electrical techniques for measuring temperature tend to average the

junction temperature, and the more nonuniform the temperature, the more the electrically

measured average temperature will deviate from the peak temperature.

Generally, it can be assumed that for chips of equal area, for the same power dissipation,

the faster the rate of cooling during measurement, the more nonuniform is the chip temperature. Also, the more nonuniform the temperature, the less representative the electrically

measured temperature is of the peak temperature.

It is possible to estimate better the peak temperature from the electrical measurement with

some further data processing. It is necessary, though, to measure the cooling response of

the device just as for extrapolating the temperature, as is shown in table 3. Because

the semiconductor chip will cool one-dimensionally for the first 200 to 300 fis, a plot of

temperature versus the square root of time will be linear during this time. The slope of the

Hnear curve is inversely proportional to the area being heated. If it is assumed also that

31
TIME (3^/2)

Figure 17. The electrically measured cooling response of a power transistor versus the

square root of time for two different operating conditions for the same power (20 W).

The infrared (IR) measured temperature is also noted for each operating condition. The

ordinate is the junction temperature rise above the reference point (case) temperature.

32
the low-level measurement current takes on the same distribution within the chip during

measurement that the heating current did during heating, then a simple relation can be

developed between the calibration curve generated by externally heating the chip (with

the measurement current uniformly distributed over the chip) and the measured TSEP

during the actual measurement (with the measurement current nonuniformly distributed over the chip). The full procedure is described elsewhere [4]. An indication of the accuracy of this technique can be seen from figure 18 which shows the

manufacturer-specified SOA, the SOA as determined using the modified electrical technique for determining the peak temperature, and the actual SOA determined using an

infrared microradiometer for a 35-W triple- diffused transistor. Also shown is the SOA

which would be determined using the standard electrical measurement technique. As has

been discussed, for high current and low voltage, the specified SOA safely limits the junction temperature below Tj^^^a;) for this device. However, as the voltage increases and the

current decreases, the current begins to constrict and the specified SOA, although derated

for second breakdown, is not derated sufficiently to account for the increased current constriction. Both the infrared-determined limit and the electrical-peak junction temperature

limit show this.

Detection of Onset of Current Constrictions - There are a number of rather simple techniques that detect when the junction temperature (or equivalently the power density distribution) begins to become nonuniform.

One possible method of detecting the onset of current nonuniformity is by monitoring Veb

at constant Ic as Vqe is increased [25]. The value of Veb can be measured either near

the end of a pulse (of 1-s duration, for instance) while the pulse is still on, or it can be

measured after the pvdse is removed from the device (~10 to 20 fis after removal of power)

with a small sampling current present. The latter is more sensitive to the onset of the

temperature nonuniformity, and the measurement technique is nearly the same as making

an RdjR measurement using the emitter-only switching technique. A plot of Veb versus VcE for several values of Ic is shown in figure 19 for the same 35-W triple-diffused device

shown previously. As the power is increased, Veb initially falls linearly with Vce because

of a rather uniform increase in junction temperature. As the power density begins to be

constricted, and the peak temperature begins to rise more rapidly, Veb falls more rapidly.

In each curve the break point from the initial linear region is the point where current

crowding has begun to occur. This measurement is very easy to perform, and each datum

point can be obtained with a single pulse of power applied to the device. An SOA curve based on the onset of current crowding can then be generated from these data.

Reasons for Measuring Temperature

There are many reasons why one desires to know the operating temperature of a power

device. Oftentimes, the specific reason for wanting to know the temperature dictates the

type of measurement to be made.

If one wants to know accurately the actual operating temperature of a device in operation,

33
5.0

< 2.0

LU

DC

DC-)O

DCO

ho

LU

_J

-J

o

o

1.0

0.5

0.2

0.1

1 1 1 1 1 1 1

1

1 1 II

Infrared

A Standard Electrical - 100% O Electrical Peak

OA — \. O ^72% -

rf>v A 33%

o\ ^19%

o\ a14%

1

1 1 1 1 1 1 1 1 8% ^1 ^ 1 |-

10 20 50 100 200

COLLECTOR-EMIHER VOLTAGE (V)

500

Figure 18. The manufacturer-specified SOA limits (solid line) and points along the measured SOA limits as determined by an infrared microradiometer, by the peak temperature

electrical technique, and by the standard electrical technique. The number beside each

standard electrical point is the percent of the total active area estimated to be dissipating

power for each operating condition.

34
LU

<

I—

o>

LU

CO<

CQ

LU

0.7

0.6

0.5

0.4

0.3

0.2

1 1 1 1

_

1 1

n o A

_ \ \0.4A

1.0 A \ \

—

1 1 1 1 1 1 1 V

20 40 60 80 100 120 140 160

COLLECTOR-EMITTER VOLTAGE (V)

Figure 19. The measured Veb at 20 /zs after the removal of a 1-s power pulse versus Vce

for a ntunber of values of Ic- The onset of current crowding is noted by the initial break

in the linear relation of Vbe versus Vce a-s Vce is increased.

35
then a careful measurement of the thermal resistance is required. This requires that dc

measurements for pulse lengths approximating many seconds be made. Also, careful heat

sinking and reference temperature measurement are required. In all, such a measurement

is long and difficult. If one truly wants to know the dc temperature, there are really no

shortcuts. Fortunately, such measurements can be made on a sampling basis on a few

devices from a lot.

Oftentimes, one wants to "compare" devices to one another. Such is the case in performing

thermal measurements for die-attach evaluation. Usually, devices of the same type differ

in thermal resistance because of differences in die-attach quality. Well-bonded chips have

a lower thermal resistance than poorly bonded chips. One can screen out poorly attached

dice by comparing the thermal response of the devices. Since it is mainly the die attach

that causes differences in the thermal resistance, it is advantageous to make the die attach

as large a part of the measurement as possible. Heat flows from the chip to the die

attach, to the heat spreader, to the package, to the heat sink. The dc thermal resistance

measurement adds all of these factors to one another. The die-attach portion is a small

part. If only the die-attach portion differs between devices, then one is looking for a small

change in a big number if the dc thermal resistance is used. It would be ideal if one could

measure only the "thermal resistance" of the die attach. Closer to practicality, but still close to ideal, it woiild be satisfactory to include only the thermal resistance of a few of the

other heat transfer regions (chip, die attach, heat spreader, package, heat sink). This can

be accomplished by heating the device with a short pulse (ranging from 10 to 100 ms for

most device types), short enough that the heat energy only has time to flow through the

chip and into the die-attach area. The pulse is stopped before significant heat enters into

the heat spreader. In this way, the measured temperature is a function of the chip and the

die attach and does not depend upon the other transfer elements. It must be emphasized,

though, that all one is testing here is the die attach. Usually, the dc thermal characteristics

follow the die-attach characteristics, but not always. Nevertheless, a pulsed measurement

used to detect the right parameter, i.e., die-attach quality, is certainly better and easier

than performing a dc test [13]. The advantages of this technique are summarized in table 4. A die- attach test, or a theoretical calculation, does not replace the need for accurate

characterization of the transient and dc thermal resistance of a device. The measurements

do not have to be made on every device, but they must be made on enough devices to assure the manufacturer and user that well-bonded devices will have a thermal resistance

less than some agreed-upon value. A true-life example of the need to know the dc thermal

resistance is in order. A device user was experiencing a high failure rate in one particular

bipolar transistor in a motor drive system. The particular device had two Darlington chips

paralleled in a single TO-3 package. An obvious question was, "How well are the chips

sharing the workload?" If one chip is poorly die bonded compared to the other, then it may end up hogging all the current due to the positive thermal feedback effects for which

bipolar transistors are well known. Pulsed tests were made out to pulses 1 s in length,

and no particular differences were noticed between devices. Also, the 1-s measurements

indicated that the dc thermal resistance was within specifications when using theoretical

thermal response curves. Careful dc thermal resistance measurements, though, revealed

36
TABLE 4

Advantages of Thermal Response Testing

for Die Attachment

The technique is:

• SENSITIVE - It can be used to detect contiguous voids as smedl as 10 to 15% of

the chip bonding area.

• CONVENIENT - It can be performed on finished devices and requires minimal, if any, heat sinking.

• FAST - It can be automated.

• ADAPTABLE - It can be used for testing a wide variety of device types and structures.

37
that there was in fact a very wide variation in the thermal resistance of different devices.

Subsequent measurements made with an infrared microradiometer revealed that in fact,

oftentimes, only one of the chips was dissipating power while its parallel mate was along

just for the ride. Qualitative infrared measurements showed that the pulsed measurements

were accurate; the chips shared power equally for single pulses at least as long as 1 s. The

problem was that because of the relative locations in the package of the two chips and the

asymmetry of the heat spreader in the TO-3 package, the thermal resistance of one of the

chips to the heat sink (the one that failed 80 percent of the time) was greater than that of

the other chip. Thus, even if each chip shared evenly in the workload, one would be hotter

(and thus eventually could not share equally because of the positive thermal feedback of

bipolar transistors). The temperature rise during a single pulse was not enough to cause

the chips not to share the pulse.

Measurement of Integrated Power Devices

The measurement of integrated power structures presents special difficulties. The major

problem is that a simple, well-defined and well-behaved TSEP often is not available at the

device terminals. Even if a TSEP is easily available, it may not be sensing the temperature

at the desired region of the chip.

For instance, consider perhaps the simplest integrated power circuit, the monolithic, bipolar Darlington transistor. A circuit schematic of a typical Darlington is shown in figure

20. The actual chip for this Darlington is shown in figures 4 and 5. The output transistor Q2 dissipates the large fraction of the power in this device and is expected to be the

hottest area on the chip. The emitter-base voltage is typically used as a TSEP for bipolar

transistors, but for the Darlington, the emitter-base voltage measured at the terminals

of the device is the sum of the emitter-base voltage of the input transistor, Ql, and the

output transistor, Q2. Under most circumstances, then, the temperature measured using

the emitter-base terminal voltage of the Darlington is some average temperature of the

input and output device. There are no readily available temperature-sensitive parameters

that one can use that only senses the output transistor temperature. This example is perhaps the simplest example of the problems associated with measuring the temperature

of integrated devices.

Although this measurement cannot be made directly, an indirect determination of the

Darlington output transistor thermal resistance using the emitter-only switching technique

and a simple equation has been developed [26]. The procedure employs two separate

measurements, one of the average temperature of the input and output transistors and

the other of the temperature of the input transistor only. The method relies upon the

presence of the two resistors, Rl and R2, and upon the fact that for very small values of

emitter-base current, only the input transistor is active. This is because the voltage drop

across R2 for very small currents may be made less than a diode forward voltage drop.

This keeps transistor Q2 off". The method is outlined in table 5. The Darlington example is indicative that perhaps simple thermal resistance measurements

may not suffice for complex integrated power circuits. Many such circuits do have isolation

38
39
TABLE 5

Procedure for Measuring the Temperature

of Darlington Transistors

(see fig. 20 for Darlington details)

•On

1. Using the emitter-base voltage of the Darlington as the TSEP, use the emitter-only

switching method and a measurement current small enough to only turn on the

input transistor (Ql) to measure the temperature of Ql. The measurement current

is selected to be small enough that the potential drop across R2 is less than about

0.3 V. Call this temperature T(l).

2. Increase the measurement current such that both transistors Ql and Q2 are turned

on during measurement. The current should be large enough that the potential

drop across R2 is greater than about 1.0 V. An average temperature of Ql and Q2

is measured with this step. Call the temperature measured T(l+2).

3. The temperature T(2) of the output transistor Q2 is calculated as T(2) = 2T(l+2)

-T(l).

40
diodes, but the diode is usually not sensing the temperature of the power elements, the

temperature of most interest.

Power Transistor Thermal Standardization Activities

Presently active in the improvement and standardization of thermal characterization techniques for power transistors are the Electronic Industries Association's JEDEC JC-25,

JC-50.1, and JC-13.1 Committees on Transistors, GaAs ReHability & Quality Standards,

and Government Liaison for Discrete Devices, respectively, and the Department of Defense

through the Defense Electronics Supply Center (see table 6 for details). Technical expertise and a wiUingness to participate actively in the development of standards are always

welcomed by the various groups involved in these efforts.

II. INTEGRATED CIRCUITS [27]

Introduction

The thermal characterization of packaged integrated circuits generally takes one of three

forms. They are computer simulations and both direct (infrared) and indirect (electrical)

measurements of the integrated circuit chip surface/junction temperature [28]. Due to the

inherent difficulties in analyzing and measuring the thermal properties of active integrated

circuit chips, an approach using specifically designed thermal test chips for evaluating new

packaging schemes is finding wide acceptance in the industry. The focus of this section

is on the thermal characterization of single-chip ceramic packages, although there is some

discussion pertaining to plastic packages and hybrids.

Thermal resistance has been used over the years as an aid to device manufacturers and

users for calculating the junction temperature of operating devices. The concept of a unique

value of thermal resistance is based upon an analogy between the electrical and thermal

properties of materials, with temperature, power dissipation, and thermal resistance being

analogous to voltage, current, and electrical resistance, respectively. The thermal resistance

{Rojr) is defined as follows:

where

RejR — thermal resistance between the junction and the reference point (usually on the

device case or in the surrounding ambient) (°C/W),

Tj,Tr — temperature of the junction and the reference point, respectively, (°C), and

Fu = power dissipation in the device (W).

In applying the electrical analogy to the thermal problem and in using eq (9), it is implicitly assumed that uniform electrical power and temperature distributions exist across

41
TABLE 6

Power Transistor Thermal

Test Method Standardization

1. "Thermal Resistance Measurements for Bipolar Transistors (Emitter-to-Base Forward Voltage, Emitter-Only Switching Method)," Method 3131.2, MIL-STD-750C

Test Methods for Semiconductor Devices, Dept. of Defense, Washington, DC.

2. "Thermal Impedance Measurements for GaAs MESFETs (Constant Current ForwardBiased Gate Voltage Method)," Method 3104, MIL-STD-750C Test Methods for

Semiconductor Devices, Dept. of Defense, Washington, DC.

3. "Thermal Impedance Measurements for Insulated Gate Bipolar Transistors (Gateto-Emitter On-Voltage Method)," Method 3103, MIL-STD-750C Test Methods for

Semiconductor Devices, Dept. of Defense, Washington, DC.

4. "Thermal Impedance Measurements for Vertical Power MOSFETs (Source-to-Drain

Diode Voltage Method)," Method 3161, MIL-STD-750C Test Methods for Semiconductor Devices, Dept. of Defense, Washington, DC.

5. "Thermal Resistance Measurements of Conduction Cooled Power Transistors," EIA

RS-313-B, Electronic Industries Association, Washington, DC.

6. "Low Frequency Power Transistors (Part 4 - Thermal Characteristics)," JEDEC

Standard No. 10, Electronic Industries Association, Washington, DC.

7. "Power MOSFET's (Chapter 4 - Thermal Characteristics)," JEDEC Standard

No. 4, Electronics Industry Association, Washington, DC.

42
the junction of the microelectronic device under study and that one- dimensional heat flow occurs from a planar heat source. The assumption is also made that the thermal properties of the materials involved are not dependent on temperature. These assumptions

imply a unique value of thermal resistance for devices of a given design and construction.

Since these assumptions are not completely valid for actual semiconductor devices, care

is needed in applying the results of eq (9) to real devices. Because both monolithic and

hybrid integrated circuits do not normally have uniform power dissipating sources, nor do

the materials from which they are made have thermal conductivities that are independent

of temperature, it is imperative that the conditions under which the device is thermally

characterized are clearly described. If this is not done, it will be almost impossible to get

accurate, reproducible, and correlatable results.

In this publication, the three techniques for thermally characterizing integrated circuits are discussed in terms of their useftdness in characterizing VLSI ceramic packages. A

major problem, though, is that one is tempted to assume that one (or possibly all) of

the techniques can be used as a standard in terms of accurately indicating the thermal

properties of packages. There is no a-priori "right" technique. At best, operating and test conditions can be defined that bear some relationship to reality and that give reproducible

results.

Design of Thermal Test Chips

A variety of thermal test chips is being used in the industry, with most companies using

chips of their own design. Despite this general use of thermal test chips by the defense

commvmity, and semiconductor chip, package, computer, and automotive companies, these

chips are still not readily available. Chips are circulated among users with little or no

documentation. A chip that is suitable for die- attachment quality- control screening may

not be usable for the thermal characterization of packages. Also, chips of the same physical

size may have very different heat-source and temperature-sensing configurations. The

implications of a particular chip design should be understood in order that the measured

results might be rationally interpreted.

Thermal test chips currently used in the industry employ planar technology and have

separate heating and temperature-sensing elements. Examples of pictorial representations

of thermal test chips are given in figure 21; actual photographs are shown in figure 22.

As can be seen, there is Httle commonality in layout and size. In all cases, p-n junctions

are used as the temperature-sensing elements, although not all of the diode elements are independently accessible (see inserts in fig. 22). Heating elements are diffused resistors

in a variety of series-parallel combinations in all but one of the examples shown in figure

21. The advantage of separate heating and sensing elements over the normal techniques

for measuring the thermal resistance of discrete semiconductor devices [12], and operating

integrated circuits [29], is that no electrical switching is involved. How then does one choose

between the various thermal test chip designs either presently in use or being proposed

for future production? One approach that has been shown to be feasible is that of using

computer simulations of the chip-package configuration to gain insight into how best to

design and utilize thermal test chips [27,30].

43
IQ 01 le

ID CI IB

Ql 10

01 I 1

100 mil

2.54 mm

I Sensing elements

Heating elements

Figure 21. Pictorial representation of heating and sensing areas on thermal test chips.

44
(b)

(a)

(d)

Figure 22. Photomicrographs of representative thermal test chips with inserts showing

sensing diode circuit configuration, (a) Signetics; (b) Motorola; (c) Burroughs; (d) Texas

Instruments.

45
Computer Simulations - Theoretical analysis for thermally characterizing semiconductor

devices has taken many forms, from the simplistic to the complex. Computer simulations

generally take one of the following forms: numerical analysis using finite element or fi-

\ nite difference codes [31,32], closed-form analytic solutions using Laplace and/or Fourier

transforms, or Green's function methods for solving the heat flow equation [33-36]. It

is the task of users of computer codes to ascertain which technique will do the required

job adequately [36,37]. For example, in a study conducted by Oettinger [27] and Albers

[30], the steady-state temperature distribution in thermal test chips bonded to various

substrate layers was simulated using the model originally developed by Kokkas [34]. The

structure modeled is depicted in figure 23 and consists of three layers with an arbitrary

number of heat sources on the surface of the top (first) layer. The rectangular dimensions

of the three layers are identical, although the thermal conductivity and thickness of each

layer may differ. The bottom of the third layer is assumed to be in contact with an ideal

heat sink and is held at a constant temperature. The model assumes only conductive heat

transfer through the three layers and no heat transfer from or through interconnection

leads. This assumption has been shown by Baxter to be valid for a wide variety of integrated circuit package configurations in which there is a preferred heat flow path, i.e., for

most ceramic packages [28]. The simplicity of the model leads to a closed-form analytic

solution which lends itself to the calculation of the temperature at any point, or points, on

or inside the three-layer structure. The computer code to implement this solution, written

in FORTRAN and designated TXYZ, has been documented by Albers [30].

Thermal Properties of Materials - One of the difficulties that occur when one attempts

to model semiconductor structures is that accurate information on the thermophysical

properties of the materials involved is not readily available. For steady-state thermal simulations, the thermal conductivity of the various materials is the property of particular

interest. The thermal conductivity of silicon and of both alumina and beryllia are very

temperature dependent [38,39]. Also, the thermal conductivity of ceramics is very dependent on the purity of the material [40]. Examples of these variations are shown in figure

24 for silicon, alumina, and beryllia. The situation is worse for die- attachment materials

such as solders, eutectics, and conductive and nonconductive epoxies. For these materials,

accurate values of thermal conductivities are hard to find. At the very least, a researcher

should indicate what values of thermal conductivity are used, and the range of temperature

and material purity for which they are valid.

Guidelines for Thermal Test Chip Design - In a study conducted by Oettinger [27] and

Albers [30], thermal test chips representative of industrial practice were modeled to determine their efficiency in adequately characterizing the thermal properties of VLSI package

structures. The structures modeled simulate a silicon chip 15 or 30 mil (0.38 or 0.76 mm)

thick, bonded with a 1-mil- (0.025-mm-) thick layer of either gold-silicon eutectic or silverfilled epoxy, to a 30-mil- (0.76-mm-) thick substrate of either alumina or beryllia. The

values of thermal conductivity for the various materials used in this study are given in

table 7. Based on the results of the computer simulations of various chip-substrate configurations, the following recommendations are made for the design of thermal test chips for

VLSI package characterization:

46
DISTRIBUTED

HEATERS

SEMICONDUCTOR

1 Lc 1 DIE ATTACH

Li t SUBSTRATE

HEAT SINK

Figure 23. Three-layer structure modeled by computer

47
vE

o

>

>

O

O

z

O

O

LU

I

TEMPERATURE (K)

E

o

>->

IO

ZD

O

z

O

O

erUJ

I

I10 0

io-ib

10 -2

10

99.5% pure

96% pure

94% pure

90% pure

1

1—I—I—

n

J \ I L_L

TEMPERATURE (K) 10

Figure 24. Influence of temperature on the thermal conductivity of integrated-circuitrelated materials. Thermal conductivity of various materials (top); thermal conductivity

of aluminum oxide for various purities (bottom).

48
TABLE 7

Thermal Conductivity of Materials

Conversion Factor: 1 cal/s = 4.184 W; k = Thermal Conductivity

Silicon

k(T) = 286/[T-100]

k = 1.05 W/cm °C

k = 0.976 W/cm °C

Copper

k = 3.93 W/cm °C

Kovar

k = 0.21 W/cm °C

Steel [1010]

k = 0.46 W/cm °C

Alumina (99-99.5%)

k = 0.25 W/cm °C

@T = 300-600 K

@T = 100°C [k = 0.00267 W/mil °C]

@T = 120°C [k = 0.00248 W/mil °C]

20-100°C [k = 0.00998 W/mil °C]

20-100°C [k = 0.000533 W/mil °C]

20-100°C [k = 0.00117 W/mil °C]

20°C [k = 0.00064 W/mil °C]

^T =

^T =

)T =

Gold-Silicon Eutectic

k = 2.95 W/cm °C [k = 0.0075 W/mil °C]

Solder

k = 0.35 W/cm °C [k = 0.0009 W/mil °C]

Epoxy (Electrically Conductive — Silver Filled)

k = 0.016 W/cm °C [k = 0.00004 W/mil °C] @120°C

Epoxy (Electrically Insulative — Alumina Filled)

k = 0.004 W/cm °C [k = 0.00001 W/mil °C] @120°C

Beryllia (98%)

k = 2.05 W/cm °C [k = 0.0052 W/mil °C] @20°C

k = 1.54 W/cm °C [k = 0.0039 W/mil °C] @100°C

Alumina (90-92%)

k = 0.17 W/cm °C [k = 0.00043 W/mil °C] @20°C

k = 0.15 W/cm °C [k = 0.00038 W/mil °C] @100°C

49
1. The heat sources, i.e., the resistor stripes or transistors, should utilize as much of

the active chip area as possible so that the measured package thermal resistance is indicative of the chip size being used. A 10-mil- (0.25-mm-) parameter stripe should

be sufficient for bonding pads, etc. Examples of the variation of chip surface-to- case thermal resistance with chip size, for a silicon chip eutectically bonded to either

alumina or beryllia substrates, are given in figure 25. It is desirable to use a range

of test chip sizes so that the package thermal resistance can be determined as a

function of chip size. Arrays of test chips can be built up from, for example, a

100-mil- (2.54-mm-) square standard chip size.

2. Spacing between heat sources, which is needed to accommodate the temperaturesensing elements, i.e., diodes, should be minimized. The spacing should be less

than or equal to 2 mil (0.051 mm). The sensing element should be located at the

center of the chip surface and should be as large as possible, consistent with design

rules. Examples of the temperature distribution of thermal test chips utilizing these

design rules are depicted in figure 26 for silicon chips eutectically bonded to either

alumina or beryllia substrates. The chips consist of nine 100-mil (2.54-mm) basic

cells, a single 300-mil- (7.62-mm-) square chip cell, and a chip with uniform heat source coverage over the entire active area (an ideal case).

3. The thermal test chip should be designed such that its power dissipation limitations are consistent with the range of package thermal resistance encountered. The ability

to cause a chip surface-to-case temperature difference of at least 20 °C is desirable.

The heat source power dissipation needed to produce a chip surface-to-case temperature difference of 20 °C is presented in figure 27, for a silicon chip eutectically

bonded to either alumina or beryllia substrates.

Following these recommendations, the computer simulations for these package configurations predict that the temperature sensor will measure a temperature that is at least 90

percent of the peak temperature on the chips. This was also found to be the case if the sensor temperature was compared to the peak temperature on the surface of an ideal thermal

test chip where the total active chip surface area was covered by a uniform heat source. The chip surface-to-case temperature difference versus distance curves, depicted in figure

28 for various test chip designs, show the temperature variations that might be encountered. It should be noted that the heat source layout depicted in chip A does not adhere

to the proposed guidelines. In general, even for chip designs that adhere to the indicated

guidelines, the higher the substrate thermal conductivity, the worse the agreement.

A comparison was made between measured results using a thermal test chip and the TXYZ

predictions for a ceramic leadless chip carrier package. The thermal resistance data, using an early Signetics thermal test chip design (see fig. 22(a)) [41] were suppHed by Gorman

[42]. The measured junction-to-case thermal resistance obtained was not indicative of

what one would expect for a chip of the particular size used. The chip heating and sensing

element layout, the measured data, and curves of computed results for the actual (curve A),

ideal (curve B), and proposed (curve C) thermal test chip designs for the 180-mil- (4.57-

mm-) square chip used in the study are presented in figure 29. The data indicate that this

50
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z

<H

(0

if)

UJ

-I

<

QC

III

LU

C/) <

oI Io

10

8

T r

Chip thickness = 15 mil

Chip thickness = 30 mil

120 160 200 240 280 320 360 400

CHIP SIZE (mils square)

T \ 1 1 r r I I I I I r

Chip thickness = 15 mil

Chip thickness = 30 mi

120 160 200 240 280 320 360 400

CHIP SIZE (mils square)

Figure 25. Influence of chip size on the junction-to-case thermal resistance of a three-layer

package structure. Alumina substrate with thickness of 30 mils (top); beryllia substrate

with thickness of 30 mils (bottom).

51
2 r I I I I I I

\ I i__j I I I I [ I \ I I I I I L 0 40 80 120 160 200 240 280

DISTANCE (mils)

Figure 26. Influence of heat source layout on chip surface temperature for test chip arrays

and single cell chips.

52
120 160 200 240 280 320 360 400

CHIP SIZE (mils square)

Figure 27. Influence of chip size on internal power dissipation for a junction-to-case temperature difference of 20 °C.

53
oo

LU

QCZJ<

UJ

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15

14

13 -

12 -

11

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10 15 20 25 30 35 40

DISTANCE (mils)

45 50 55 60

Figure 28. Influence of heat source layout on chip surface temperature for a silicon chip

eutectically bonded to an alumina substrate.

54
DISTANCE (mils)

Figure 29. Comparison of computer-simulated and measured thermal resistance for a

leadless ceramic chip carrier package.

55
particular thermal test chip design would give overly conservative results in characterizing

the thermal properties of a similar package with the same size chip with uniform power

dissipation. The quantitative nature of the TXYZ code is also demonstrated.

Fabrication techniques for implementing thermal test chips vary and are not limited to

bipolar technologies. Table 8 lists a number of possible techniques for fabricating thermal

test chips using both bipolar and MOS technologies. Most of these approaches have been

utilized at one time or another.

Utilization of Thermal Test Chips

Assuming that one is satisfied with the design of a thermal test chip and is ready to

measure the thermal characteristics of a particular chip package configuration, then both

direct and indirect procedures, along with a variety of mounting arrangements, are available

for measuring the chip-surface/junction temperature of the packaged device.

Mounting Considerations - At present, there is no consensus as to the preferred mounting

arrangement for thermally characterizing VLSI packages. There is even difficulty in determining how to quantify what is "best." Also, when one discusses mounting arrangements,

it becomes apparent that the point at which the reference temperature is measured is an important consideration. Should junction-to-case or junction-to-ambient temperature

measurements be made? At what specific location on the device case or in the surrounding

ambient environment should the reference point temperature be measured?

There are four basic mounting configurations used when thermally characterizing packaged

microelectronic chips. These mounting methods, pictorially depicted in figure 30, potentially allow for measurements of both case and ambient environment temperatures. All of

these mounting methods are presently being used to characterize thermally VLSI packages,

although there appears to be little agreement between the various approaches. Baxter, in

his studies to identify a preferred technique for thermally characterizing military grade

integrated circuit packages [28], indicates that using a reference point on the package for

junction-to-case thermal resistance measurements can lead to reproducible results that are independent of the type of package heat-sinking used. For packages with a preferred heat

flow path, the recommended case reference point is the point of majcimum temperature

on the package. This reference point is determined with the device operated in free air

and with no external heat-sinking. For ceramic and metal packages, this point is generally

located on the package surface directly below the chip. It should not be assumed, though,

that all devices have a preferred heat flow path. It is unclear at this time whether this

concept can or should be applied, for example, to plastic-encapstdated devices. In fact, in a study conducted by Hannemann [43], in which plastic-encapsulated devices were included, an approach using the ambient as the reference point for microelectronic device thermal

resistance measurements was advocated.

Computer simiilations in support of the concept of a measured thermal resistance that is independent of heat-sinking were reported by Bajcter et al. [44], and are shown in table 9

where computed junction-to-case thermal resistance {Rqjc) data are shown to be constant

56
TABLE 8

Fabrication Technologies for Thermal Chips

MOS Technology

Polysilicon layer for heating and drain/source junctions for temperature sensing

Transistors with the gate shorted to the drain for heating and p/n junctions for

temperature sensing

Discrete power transistor for heating and for temperature sensing

Power transistors for heating and p/n junctions for temperature sensing

Bipolar Technology

Diffused resistors for heating and p/n junctions for temperature sensing

Discrete power transistor for heating and for temperature sensing

Buried layer for heating and junction diodes for temperature sensing in an EPI

layer immediately above

Power transistors for heating and p/n junctions for temperature sensing

57
Fluid Bath Heat Sink

Wind Tunnel Still Air

i

Figure 30. Package mounting arrangements.

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59
for a variety of heat-sinking conditions. Support of this concept is also seen in figure 31

in which Brown [45] plots measured thermal impedance curves for a ceramic package with an integral aluminum heat sink under a variety of mounting and heat-sinking conditions.

Note that for times less than approximately 10 s, the thermal impedance for all mounting

conditions is the same. Although these results are encouraging, further studies are needed

since agreement between mounting techniques such as the temperature-controlled heat sink

and the fluid bath is generally poor.

The temperature-controlled heat-sink mounting method, which is usable for thermally

characterizing packaged devices for both the direct and indirect measuring methods, is the

most widely documented approach [12,46]. An example of a temperature-controlled heat

sink for a dual-in-line package is shown in figure 32. In general, detailed documentation

for the mounting methods depicted in figure 30 does not exist in the open Hterature.

Direct Measurement Procedures - Techniques for measuring the temperature of semiconductor chip surfaces have been utilized for many years. Approaches using liquid crystals,

thermographic phosphors, laser scanners, liquid-gas (bubble) formation techniques, and

infrared microradiometers have been quantified and used with varying degrees of success

[7,9-11,47-51]. Each of these techniques presents a unique set of problems ranging from

cost effectiveness and spatial and temperature resolution to ease of implementation and

degree of contamination. The techniques are generally considered to be nondestructive,

although this may be a matter of semantics since in all cases the chip surface is exposed

and in most cases it is coated with a substance that is a potential contaminant. Since

the infrared technique is the most widely used, it is the only one considered here, and the

information presented has some relevance to the other techniques.

A pictorial representation of an infrared microradiometer is shown in figure 33. For

semiconductor device studies, the temperature-sensing element used is generally a liquidnitrogen-cooled indium antimonide detector. A good discussion on the use of infrared

techniques for measuring the temperature of integrated circuits can be found in two papers published in the late sixties [50,51]. The information presented in table 10 summarizes

some of the precision and accuracy considerations to be addressed [51], even today, if one

is to utilize effectively the infrared technique.

Indirect Measurement Procedures - Temperature-sensitive electrical parameters of the particular packaged device being characterized are used as indirect indicators of chip sur- face/junction temperature in this measurement procedure. Generally, the forward voltage

of a p-n junction is used as the temperature-sensitive electrical parameter. Resistors,

diodes, or transistors are used to generate heat at or near the chip surface. The case temperature is measured on the outside of the package at a point in the main heat flow

path between the chip and heat-dissipating medium. For discrete semiconductor devices,

the temperature-sensing element is not separated from the heating element as is the case

for most presently used integrated circuit thermal test chips. When the heating element

also incorporates the temperature-sensing element, a switching method is used in implementing the indirect measurement procedure [46]. When the heating and sensing elements

are separate, no switching is involved. Thermal test chips can be designed in which the

60
0.1 0.5 1.0 5.0 10 20 30

TIME (min)

Figure 31. Thermal response measurements for an alumina chip carrier package with an

integral aluminum heat sink for various mounting and cooling configurations.

61
Figure 32. Example of a temperature-controlled heat sink.

62
Figure 33. Example of an infrared microradiometer. (After Bajcter et al., Ref. [44])

63
TABLE 10

recision and Accuracy of Infrared Measurements

Hardware-Dependent Factors

• Mechanical instabilities

• Changes in chopper temperature

• Optical defocusing due to temperature changes

• Specimen ambient temperature fluctuations

Calibration Considerations

• Variations in specimen emissivity

• Variations in specimen transmission

• Poor constant emissive coatings

• Point-by-point calibrations for thermal mapping

64
sensing element is either separate from or incorporated in the heating element. Also, if the

heating element is a transistor, care should be taken to avoid operating conditions where

current crowding occurs [2,52]. To calculate the packaged device surface/junction-to-case

thermal resistance, the following equation is used:

^

{Tj - Tc) ^ {Vmh - Vmc) ( AVmc

'''' P{Avg) Ph \ATmc

-1

(10)

Im'

wnere

Im = measuring/calibration current (mA);

Vmh = value of temperature-sensitive parameter (mV), measured at Im and

corresponding to the temperature of the chip surface/junction heated

by Ph;

Tmc — calibration temperature (°C), measured at the reference point on the

device case; Vmc — value of the temperature-sensitive parameter (mV), measured at Im and

a specific value of Tmc\ and

AVmcI ATmc = temperature coefficient of the temperature-sensitive parameter (mV/°C),

measured at ImEquation (10) is used to calculate thermal resistance for both switched and nonswitched

chip designs. The general procedure for implementing the measurement can be summarized

as follows [46,53]. The temperature coefficient of the temperature-sensitive parameter is obtained by externally heating the device in an oven or on a temperature-controlled heat

sink. The temperature-sensitive parameter is measured as a function of the reference point

temperature at a specified constant measuring current. The reference point temperature

range used during this calibration procedure should include the temperature range en- countered when internal heating power is applied to the device. The measuring current

is generally chosen such that the temperature-sensitive parameter decreases linearly with

increasing temperature over the range of interest and that negligible internal heating occurs. The value of the temperature coefficient of the temperature-sensitive parameter is then calculated from the calibration curve, Vmc versus TmcWhen internal heating power is applied to the device, the reference point temperature is held constant at a preset value. First, Vmc is measured with the same measuring current

used during calibration. Then the device is operated with steady-state heating power

applied. Vmh is then measured, with the same measuring current used during calibration.

P-eJC can then be calculated using eq (10).

Details for implementing the indirect techniques for measuring thermal resistance can be

found in the literature [12,29,46,53] and are not discussed here. Some general comments

65
are in order, however. In implementing this type of measurement procedure, the following concerns should be addressed:

1. Long-term, single-operator, repeatability measurements of junction-to-case thermal

resistance of power transistors, mounted on a temperature-controlled heat sink,

had a standard deviation of the resulting junction-to-case temperature difference

of approximately 0.5 °C [12]. It is for this reason that a chip surface/junction-toreference-point temperature difference of at least 20 °C is desirable. However, a

potential difficulty with producing a large temperature difference is that there is a tendency to exceed the power dissipation design rating of the integrated circuit or thermal test chip being measured. Although the device may not be destroyed

or degraded, what sometimes occurs is that the IR drop in the leads and contacts

becomes significant, which can lead to an overestimate of the power dissipation in

the active area of the device.

2. The temperature at a specific location on the case of the device being thermally

characterized is held constant at a preset value when internal heating power is applied. Because the temperature of the device package is not uniform, it is important

to be able to measure at all times the temperature at the same location on the package. While the device is dissipating internal power, the temperature gradient along

the package surface which is in contact with a temperature-controlled heat sink is a function of the package material thermal conductivity. For example, for a TO-3

encased power transistor at 60 W of power dissipation, a copper case can have a temperature difference from an extreme edge of the bottom to a point on the bottom

directly beneath the chip (a distance of about 2 cm) of about 2 °C; an aluminum

package, a difference of about 3 °C; and a kovar or steel package, a temperature

difference of about 6 °C [22]. Because most integrated circuit packages are made

with materials which have a thermal conductivity that is lower than kovar, it is very

important to measure the temperature each time at the same location on the case.

IC Thermal Standardization Activities

Standardized measurement techniques to thermally characterize integrated circuits have

been of interest to the semiconductor industry and its customers for many years [29]. However, these techniques come from the body of work that was specifically directed towards

conduction- cooled power transistors. Little work had been focused specifically on still- and

forced-air convective cooling of integrated circuits. Until recently, the impetus for standardized measurement techniques in this area came from the requirement for MIL-SPEC parts

[28,46], and the focus has generally been on ceramic packages that were conduction-cooled.

Recently, the computer industry has taken a more active role in efforts to provide standardized techniques for the thermal characterization of integrated circuits. The computer

industry is generally concerned with plastic-encapsulated and ceramic-packaged devices

that are, for the most part, convection cooled. Thus, the present concern in some quarters

is for standardization of package-mounting arrangements and wind tunnels [43,54-56].

Presently active in the improvement and standardization of thermal characterization tech66
TABLE 11

IC Thermal Test Method and Specification Standardization

1. "Accepted Practices for Making Microelectronic Device Thermal Characteristics

Test - A User's Guide," JEDEC Engrg. Bull. No. 20, Electronic Industries Assoc.,

Washington, DC.

2. "Thermal Characteristics," Method 1012.1, MIL-STD-883C Test Methods and Procedures for Microelectronics, Dept. of Defense, Washington, DC.

3. "Unencapsulated Thermal Test Chip," SEMI G32-86 GuideHne, 1989 Book of SEMI

Standards, Vol. 4, Packaging Div., SEMI, Inc., Mountain View, CA.

4. "Junction-to-Case Thermal Resistance Measurements of Molded-Plastic Packages,"

SEMI G43-87 Test Method, 1989 Book of SEMI Standards, Vol. 4, Packaging Div.,

SEMI, Inc., Mountain View, CA.

5. "Still- and Forced-Air Junction-to-Ambient Thermal Resistance Measurements of

IC Packages," SEMI G38-87 Test Method, 1989 Book of SEMI Standards, Vol. 4,

Packaging Div., SEMI, Inc., Mountain View, CA.

6. "Junction-to-Case Thermal Resistance Measurements of Ceramic Packages," SEMI

G30-88 Test Method, 1989 Book of SEMI Standards, Vol. 4, Packaging Div., SEMI,

Inc., Mountain View, CA.

7. "Thermal Test Board Standardization for Measuring Junction-to-Ambient Thermal

Resistance Measurements of Semiconductor Packages," SEMI G42-88 Specification,

1989 Book of SEMI Standards, Vol. 4, Packaging Div., SEMI, Inc., Mountain View,

CA.

8. "Thermal Transient Testing for Die Attachment Evaluation of Integrated Circuits,"

SEMI G46-88 Test Method, 1989 Book of SEMI Standards, Vol. 4, Packaging Div.,

SEMI, Inc., Mountain View, CA.

67
niques for integrated circuit chips and packages are the Electronic Industries Association's

JEDEC JC-13.2 Committee on Government Liaison for Microelectronic Devices, the Department of Defense through the Defense Electronics Supply Center, and the Semiconductor Equipment and Materials International's (SEMI) Ceramic and Plastic Packages

Standards Committees (see table 11 for details). Technical expertise and a willingness to

actively participate in the development of standards are always welcomed by the various

groups involved in these efforts.

Acknowledgments

The authors wish to thank David W. Berning, Sherwin Rubin, and Robert L. Gladhill

who, over the years, have contributed to the technical work described in this publication

as indicated by the various references. We would like to thank Jane Walters, who as an

editor is without peer, for putting it all together.

The material in this Special Publication was given as a tutorial at the 1989 IEEE International Reliability Physics Symposium, Phoeniz, Arizona on April 10, 1989, and appears

in their Tutorial Notes 1989.

References

Excerpted in part from D. L. Blackburn, A Review of Thermal Characterization of

Power Transistors, Proceedings Fourth IEEE Semiconductor Thermal and Temperature Measurement Symposium, San Diego, CA, pp. 1-7, 1988.

Excerpted in part from F. F. Oettinger, D. L. Blackburn, and S. Rubin, Thermal

Characterization of Power Transistors, IEEE Trans. Electron Devices ED-23, 831-

838 (1976).

B. Reich and E. B. Hakim, Hot Spot Thermal Resistance in Transistors, IEEE

Trans. Electron Devices ED-16, 166-170 (1969).

D. L. Blackburn, An Electrical Technique for the Measurement of the Peak Junction Temperature of Power Transistors, IEEE 13th Annual Proceedings, Reliability

Physics 1975, Las Vegas, NV, pp. 142-150, 1975.

F. F. Oettinger and S. Rubin, The Use of Current Gain as an Indicator for the

Formation of Hot Spots Due to Current Crowding in Power Transistors, IEEE 10th

Annual Proceedings, Reliability Physics 1972, pp. 12-18, 1972.

S. P. Gaur, D. H. Novon, and R. W. Teerlinck, Transistor Design and Thermal

Stability, IEEE Trans. Electron Devices ED-20, 527-534 (1973).

D.J. Brenner, A Technique for Measuring the Surface Temperature of Transistors

by Means of Fluorescent Phosphors, NBS Tech. Note 591 (July 1971).

P. L. Hower and P. K. Govil, Comparison of One- and Two-Dimensional Models

68
of Transistor Thermal Instability, IEEE Trans. Electron Devices ED-21, 617-623

(1974).

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72
NIST-114A U.S. DEPARTMENT OF COMMERCE 1. PUBUCATION OR REPORT NUMBER

(REV. 3-«9) NATIONAL INSTITUTE OF STANDARDS AND TECHNOLOGY NT<;T/"^p-dnn/?!fi

2. PERF^ORMINQ ORQANIZATION REPORT NUMBER BIBLIOGRAPHIC DATA SHEET X PUBUCATION DATE July 1990

4. TTTLE AND SUBTTIXE Semiconduatov Measurement Technology: Thermal Resistance Measurements

5. AUTHOR(S)

F. F. Oettinger and D. L. Blac3cbiirn

6. PERFORMINQ ORQANIZATION (IF JOINT OR OTHER THAN NIST, SEE INSTRUCTIONS)

U.S. DEPARTMENT OF COMMERCE

NATIONAL INSTTTUTE OF STANDARDS AND TECHNOLOGY

aArrHEr;SBURa, md 20899

7. CONTRACT/QRANT NUMBER

8. TYPE OF REPORT AND PERIOD COVERED Final

9. SPONSORINQ ORQANIZATION NAME AND COMPLETE ADDRESS (STREET, CITY, STATE, ZIP)

Same as item #6

10. SUPPLEMENTARY NOTES

DOCUMENT DESCRIBES A COMPUTER PROGRAM; SF-185, F1P8 SOFTWARE SUMMARY, IS ATTACHED.

11. ABSTRACT (A 200-WORO OR LESS FACTUAL SUMMARY OF MOST SIGNIFICANT INFORMATION. IF DOCUMENT INCLUDES A SIGNIFICANT BIBUOQRAPHY OR

LTTERATURE SURVEY, MENTION IT HERE.)

Tius Special Publication reviews the thermal properties of power transistors ajid integrated circuits and discusses methods for chaxacterizing these properties. The discrete

devices discussed include bipolax trajisistors and metal-oxide-semiconductor field-effecttransistors. Measurement problems common to these devices, such as deciding the reason

for requiring a paxticular measurement, adequate reference temperature control, selection

of a temperature-sensitive electrical parameter, and separation of electrical and thermal

effects during measurement axe addressed. Due to the inherent difficulties in measuring and analyzing the thermal properties of active integrated circuits, an approach using

specifically designed thermal test chips for evaluation of new die-attachment and packaging

schemes is finding wide acceptance in the industry. In this Special Publication, indirect

(i.e., electrical) measurements, direct (e.g., infrared) measurements, and computer simulation techniques for thermally characterizing integrated circuits are discussed in terms of

their usefulness in chaxacterizing VLSI packages.

12. KEY WORDS (6 TO 12 ENTRIES: ALPHABETICAL ORDER: CAPITAUZE ONLY PROPER NAMES; AND SEPARATE KEY WORDS BY SEMICOLONS)

integrated circuits; junction temperature; power transistors; safe-operating-area

limits; temperature-sensitive electrical parameter; thermal measurements; thermal

resistance; thermal test chips

13. AVAILABIUTY UNUMITED

FOR OFFICIAL DISTRIBUTION. DO NOT RELEASE TO NATIONAL TECHNICAL INFORMATION SERVICE (NTIS).

XX

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ORDER FROM SUPERINTENDENT OF DOCUMENTS, U.S. GOVERNMENT PRINTING OFFICE, WASHINGTON, DC 20402. ORDER FROM NATIONAL TECHNICAL INFORMATION SERVICE (NTIS), SPRINGFIELD. VA 22161.

14. NUMBER OF PRINTED PAGES

78

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