NBSIR 75-805

CALIBRATION TECHNIQUES FOR ELECTROMAGNETIC

HAZARD METERS: 500 MHz to 20 GHz

Ronald R. Bowman

Electromagnetics Division

Institute for Basic Standards

National Bureau of Standards

Boulder, Colorado 80302

April 1976

Prepared for

Calibration Coordination Group

Army/Navy /Air Force

NBSIR 75-805

CALIBRATION TECHNIQUES FOR ELECTROMAGNETIC

HAZARD METERS: 500 MHz to 20 GHz

Ronald R. Bowman

Electromagnetics Division

Institute for Basic Standards

National Bureau of Standards

Boulder, Colorado 80302

April 1976

Prepared for

Calibration Coordination Group

Army /Navy /Air Force

U.S. DEPARTMENT OF COMMERCE, Elliot L. Richardson, Secretary

James A. Baker, III, Under Secretary

Dr. Betsy Ancker-Johnson , Assistant Secretary for Science and Technology

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Acting Director

CONTENTS

Page

1. INTRODUCTION 1 1.1 Objectives 1 1.2 Orientation 2 1.3 Calibration Methods Considered 2 2. PROBLEM DEFINITION 3 2.1 Introduction 3 2.2 The Proximity Correction Method: Some Relevant Discussion 3 2.3 The Importance of Antenna Aperture Size for Electromagnetic

Hazard Meter Calibrations 6 2.4 Some Conclusions 9 2.5 Open-Ended Hollow Waveguide Radiators 10

2.6 Summary of Experimental Work Performed 12

3. EXPERIMENTAL WORK PERFORMED 12

3.1 Gain Measurements 12

3.1.1 Introduction 12

5.1.2 Equipment Used and System Calibration 13

3.1.3 Insertion Loss Versus Distance 14

3.1.4 Proximity Corrections 16

3.1.5 Mismatch Factor 17

5.1.6 Results and Discussion 18

3.2 Near-Field Corrections 18

3.3 Applying the Results 19

3.5.1 Summary of the Calibration Method 19

3.3.2 Precautions and Suggestions 21

3.3.3 Test Calibrations 22

4. REFERENCES 23

iii
LIST OF TABLES

Page

Table 1. Measured reflection coefficients of WR430 open-ended guide,

58.8 ± 0.1 cm long -. 24

Table 2. Gain formulas for various waveguides 24

Table 3. Proximity corrections for various waveguides 25

Table 4. Net power P i0 required to establish 10 mW/cm 2 at two distances on- axis in front of open-ended WR 430 waveguide. Also shown are

the (numeric) gain G, near-field reduction 1/C, and the nearfield gain G/C 26

Table 5. Net power Pio required to establish 10 mW/cm 2 at two distances

on-axis in front of open-ended WR 650 waveguide. Also shown are

the (numeric) gain G, near-field reduction 1/C, and the nearfield gain G/C 27

Table 6. Net Power Pi 0 required to establish 10 mW/cm 2 at two distances

on-axis in front of open-ended WR 975 waveguide. Also shown are

the (numeric) gain G, near-field reduction 1/C, and the nearfield gain G/C 28

Table 7. Net power P 10 required to establish 10 mW/cm 2 at two distances

on-axis in front of open-ended WR 1500 waveguide. Also shown are

the (numeric) gain G, near-field reduction 1/C, and the nearfield gain G/C 29

LIST OF FIGURES

Page

Figure 1. Block diagram of equipment used for the measurement of the gain

of open-ended WR430 waveguide 30-

Figure 2. Antenna range configuration showing the important sources of

scattering. The walls and ceiling were also covered with

radiation absorbing material 31

Figure 3. Typical set of recordings of relative received power versus separation distance. The two heavy lines are drawn through the

extremit ies of the multipath interference perturbations and the

dashed line is the median between the heavy lines 32

Figure 4. Graphical determination of 1/N. The values of 1/N are obtained

by dividing the ordinates of the curve by the intercept at

1/r = 0 33

Figure 5. N versus frequency for 24 cm separation 33

iv
CALIBRATION TECHNIQUES FOR ELECTROMAGNETIC HAZARD METERS: 500 MHz TO 20 GHz

Ronald R. Bowman

ABSTRACT

The calibration techniques discussed are suitable for producing

fields for calibrating most electromagnetic (EM) hazard meters to

within ± 1.0 dB using a minimum of laboratory space and oscillator

power. Above about 2.6 GHz, adequate equipment and standards have

been available for these calibrations. Below this frequency the

large apertures of the usual horn radiators require more power than

is available from medium power oscillators. Further, calibrations

in closed systems are difficult except at frequencies well below

1 GHz. Thus there is a need for small -aperture gain standards from

about 500 MHz to 2.6 GHz. The main portion of the work reported

here consists of accurate gain measurements for open-ended hollow

waveguide radiators (OEG) for use from 500 MHz to 2.6 GHz. Other

characteristics of this type of radiator important for EM hazard

meter calibrations were also determined: near-field corrections,

reflection coefficients, and aperture scattering. The suitability

of the calibration scheme was tested by performing calibrations at

2 GHz on an EM hazard meter with both a horn radiator and an OEG

radiator

.

Key words: Calibrations; electromagnetic hazards; field meters;

gain; microwave; near-field.

1. INTRODUCTION

This report describes work performed to provide improved capability for

calibrating electromagnetic hazard meters in the frequency range from, primarily,

500 MHz to 2.6 GHz. The emphasis is on calibrations of moderate accuracy with

a minimum of laboratory space and oscillator power required. The calibration

techniques described are also suitable for performing calibrations up to 20 GHz

and beyond. The work was sponsored by the Calibration Coordination Group

of the Department of Defense.

1 . 1 Ob j ect ives

The contractual objectives of the effort were to determine satisfactory

techniques for calibrating electromagnetic (EM) hazard meters to within

± 1.0 dB from 0.9 GHz (or lower) to 20 GHz (or higher) and to perform the

necessary work to implement these techniques. Because of limited funding, it was necessary to carefully restrict the work to the most important essentials,

as discussed in section 2. The calibration techniques are also useful below

0.9 GHz, but their implementation requires increasing anechoic chamber size

and oscillator power with decreasing frequency.
1.2 Orientation

The objectives of this effort are to be accomplished within several con- straints. The calibration scheme is to use only commercially available

equipment, allow calibration at 10 mW/cm 2 or higher field levels (above

0.9 GHz), use a small amount of space, and require only moderate -power oscillators (see section 2.4).

An important consequence of these restrictions is that the volume of

the calibration field will be necessarily small. For all existing EM hazard

meters, this means that only the field sensor of the meter can be placed in

the region of (approximately) uniform calibration field and that the other

parts of the meter may be subjected to much lower field levels. Thus, for

example, an interaction between the field sensor and the case of the instrument may be quite different in the calibration situation than in a true farfield plane-wave situation. The effects of the operator and case will usually

have to be determined using larger spaces and source powers of 100 watts or more. In using the meter, these effects may or may not be important depending

on the instrument design, the field being measured, and the method of operation of the meter. The associated perturbations of the meter reading may

well be more than the desired uncertainty limits of ± 1 dB „ particularly

for the newer instrument designs that usually have omnidirectional or nearly

isotropic response. Because these effects are so variable, it is considered

desirable to minimize them while performing calibrations of the field sensor.

The scheme described here helps to provide this minimization.

1 . 3 Calibration Methods Considered

Due to limited funding, only established calibration methods were practical to implement. The proximity correction method* (see section 2.2) developed

earlier [1,2] for calibrating field intensity meters and EM hazard meters is

basically satisfactory, but some further work was necessary to implement

this method for use below 2.6 GHz. This method, as further developed, has the

following important features: (a) it requires much less oscillator power

than when using large antenna separations; (b) the calibration techniques

are suitable for small indoor antenna ranges; and (c) the techniques are

simple, involve basically the same procedure throughout the frequency range

of interest, and employ only commercially available equipment.

*The term "proximity correction" as used here and in [3,4] is identical to the

term "near-zone correction" as used in [1,2], "Proximity" has been used

more often in the open literature than "near-zone," and the author prefers

the former expression. These expressions have been defined to emphasize that

both the receiving antenna and the transmitting antenna determine in an inseparable way the power transfer between antennas that are "proximate."

In contrast, the term "near-field" is usually defined [5, p. 189;6;7] with

respect to a single transmitting antenna and is not strictly relevant for

real measurement situations where a receiving antenna or probe is always

present.

2
Some attention was given to two other alternatives:

(1) A closed system (for example, a hollow waveguide or a "TEM cell"

[8] terminated in either a short or a matched load) can be used to establish

accurately known fields. Under certain conditions probes and instruments can

be introduced into such a closed system and accurately calibrated. This

method is attractive because it requires a minimum of oscillator power and

laboratory space. However, this method is not suitable for the frequency

range of interest because only a few commercially available EM hazard meters

are capable of being calibrated by this method except at frequencies much

below 0.9 GHz (primarily because they have field probes that are too large

to be introduced into closed systems without seriously perturbing the known

field) . For a discussion of the use of this method for frequencies much

below 0 . 9 GHz , see [8] . (2) It appears feasible to develop stable, isotropic field sensors

for use as calibrated standards at frequencies at least as high as 10 GHz.

Probably, only one field probe would be required to cover the range from

10 MHz to 10 GHz. Using this method, EM hazard meters would be calibrated

simply by comparing them in the same fields (either in open space or in closed

systems) with a stable laboratory- qual ity field meter. This method could not

be implemented because the development cost is too great at the present time;

but, eventually, it may provide the simplest and cheapest calibration scheme

over a large frequency range [9].

2. PROBLEM DEFINITION

2 . 1 Introduct ion

A significant part of the work described here was to clearly determine

the further work necessary to establish ± 1.0 dB calibration capability over

at least the frequency range 0.9 to 20 GHz. By carefully restricting the

work to the most essential needs, it was possible to provide an expanded

data base so that reasonably satisfactory calibrations can be performed

with available equipment from about 0.5 MHz to over 20 GHz. The following

analysis of the calibration problem is important for understanding the present

work as well as for indicating some remaining weaknesses in the present

calibration methods for EM hazard meters.

2 . 2 The Proximity Correction Method: Some Relevant Discussion

The proximity (or "near- zone") correction method developed earlier

[1,2] for calibrating field strength meters and EM hazard meters has proven

quite satisfactory. This method is currently used by the National Bureau of

Standards and is fairly widely used elsewhere. The method uses the same

3
procedures that would be used for large separations except that small separations are used and a correction is employed to remove the associated error. Given the necessary corrections, the proximity correction method avoids the

serious difficulties (large multipath interference and much greater source power)

involved with large separation distances. Advanced techniques are available

[4] for measuring proximity corrections without using large separation

distances. Though the measurement of these corrections is sometimes difficult, the effort is justified if more than a few probes or antennas of a given type are to be eventually calibrated. Except for a few subtle and

troublesome details that are discussed below, the application of the proximity

correction method is simple.

The calibrations under discussion here are for the response of the field sensor in a plane-wave field of power density S. If large separation

distances d are used, S can be established with a known antenna of gain G by the basic formula

GP

(d - ») (2.1)

4 TTd '

where P is the net power delivered to the antenna. (It is assumed throughout

that the field point is on the axis of the transmitting antenna so that "G" means the maximum power gain.) Then if R is the reading of the meter, the

calibration can be thought of as determining the calibration constant K by

K = — = —— (d * -) . (2.2)

R 4TTd 2 R

If short separation distances are used and C is the proper proximity correction, K is determined by

K = — = ——— (2.3)

CR 4iTd 2 CR

where S is calculated from eq. (2.1) despite small d. Note that eq. (2.2) is

inherently approximate since it holds only asymptotically; but eq. (2.3) is

exact insofar as all of the quantities are known. C is determined by the characteristics of the receiving antenna or sensor as well as the characteristics of

the transmitting antenna. Further, C cannot be factored into two functions,

one dependent only on the characteristics of the receiving antenna or probe and

the other dependent only on the characteristics of the transmitting antenna [10].

Logically, therefore, C should be thought of as an overall correction applied

to the response or reading R.

4
It is tempting (particularly when the field sensor is small) to identify

G/C as the "near-field gain"* of the transmitting antenna and S/C as the

actual near-field power density. However, for real situations this identification can be correct only in an approximate sense and can be significantly

in error even when the receiving antenna or probe is small. For all existing

probes and antennas, the measurement of power density is dependent, implicitly,

on the known plane-wave relations between the electric and magnetic fields.

Strictly speaking, Maxwell's equations require that a plane-vave must be

infinite in extent and have uniform power density. For practical purposes it

is only necessary that the field in the vicinity of the probe or receiving

antenna, i.e., the "local" field, not differ significantly from the "true"

plane-wave configuration. In a plane-wave, the electric and magnetic vectors

are orthogonal to each other and to the propagation direction. Further, their

electrical phase differs by 180 degrees and their ratio is (u /e

q

)

2

. If a probe is located close to a radiator, one or more of these conditions can be

violated to a significant degree even if the local field is uniform in con- figuration. Though it may be possible to build a probe that would measure

S = | EXH* | for any field configuration, no existing probes can do this. An

accurate proximity correction includes the difference in the probe response due

to any significant differences between the far-field and near-field configurations. Though there is little direct evidence at present to substantiate the

claim, it is very unlikely that, for the on-axis separations discussed in this

report, the near- field configuration differs significantly from the far-field

configuration. (By "significant" it is meant in this discussion that the

associated error in a given measurement is more than 0.03 dB . ) Insofar as this claim is true, the proximity corrections for all small probes will be the

same for a given radiator and G/C will be essentially the near-field gain of

the radiator.

Except for the objections raised in the last paragraph, the "near-field

gain" concept is useful (and will be used in this report) . Often G/C will be

nearly equal to the near- field gain of the transmitting antenna. When the

field sensor or receiving antenna is small with respect to the local field

variations and has the same response coefficient for the near-field power

density as for the far-field power density, the ratio of the far-field gain

to the near-field gain is the proper proximity correction. For such cases,

it is easy to determine proximity corrections for many antennas by using near- field gain calculations as given, for example, in [6]. However, it is difficult

*The reader is cautioned that several definitions for near-field gain have

appeared in the literature, usually in an implicit way. Further, some of

these definitions are not physically very meaningful. As used here, "nearfield gain" is defined by eq. (2.1) except that d is finite and S is the power

density at the near-field point. (Some near-field definitions are for the

electric field strength or intensity rather than for S.)

5
to estimate the errors associated with these calculations as well as the errors associated with the above assumptions about the receiving system.

Therefore, it is considered preferable to use measured proximity corrections

even when calculated corrections are feasible.

It is important that eq. (2.3) not be misused by utilizing separations

that are too small. If d is too small, the proximity corrections may have

a significant dependence on the angular orientation of the field sensor at

the field point (a serious complication only if the rotational response of

the sensor is to be determined as part of the calibration). Also, for some types of EM hazard meters, the corrections may be a function of field intensity

level as well. This last statement may seem surprising; but it follows simply

from the fact that, unlike most antenna-receiver systems, some EM hazard probes

have energy detectors spread out over a fairly large region. For example, a popular probe utilizes a series of thin-film thermocouples distributed on lossy

strips extending about two to three centimeters. Under extreme near-field

conditions, the field may not be uniform over the dimensions of the field sensor so that the thermocouples may have a different temperature configuration

than they would have for a locally uniform field. The resulting error depends

on, among other things, the linear dynamic range of the thermocouple detectors

and could perhaps be significant under extreme near-field conditions. Obviously,

the problems discussed in this paragraph need to be investigated. In the

meantime, it is recommended that the near -zone correction method not be used

for separations less than roughly a

2 /A where a is the diameter of a circular

aperture, the side of a square aperture, or the larger side of a rectangular

aperture. This criterion may well be too conservative, particularly for some antenna-probe situations, but the use of distances much smaller than a

2 /A

should be accompanied by justification.

2 . 3 The Importance of Antenna Aperture Size for Electromagnetic

Hazard Meter Calibrations

From eq . (2.3)

KR

P 4Trd 2 C (2.4)

where KR/P is the "apparent" (as read by the meter) power density* per watt

of net power delivered to the antenna. From the fundamental relation

^ , (2.5)

A

2

*Insofar as G/C is equal to the near-field gain, KR = S/C is equal to the true

near-field power density. However, it is immaterial to the discussion in this

section whether or not KR is the true near-field power density.

6
eq. (2.4) becomes

KR A

P A

2 d2 C

where A is the effective receiving area of the antenna. Expressing the

separation distance in terms of the usual "normalizing" factor a

2 / X , KR . A

(2.6)

P r

2 a"C

(2.7)

where a is the largest dimension of the aperture and r = d(a 2 /A) . Assuming

a rectangular aperture of dimensions a and b and using the aperture efficiency

n = A/Ap where A

p

= ab is the physical aperture area, eq. (2.7) becomes

KR

P a

J r

2 CA

p Insofar as the quantities in eq. (2.8) are known, it is an exact relation.

Some rather crude approximations and assumptions will now be introduced to

provide a simple interpretation of eq. (2.8) and to estimate the approximate

oscillator power required to establish a given apparent power density in

terms only of the physical aperture area. Except for apertures with one or more dimensions of about a wavelength

or less, antennas with the same configuration of aperture illumination tend

to have roughly the same aperture efficiency regardless of the aperture

size. Further, for a given value of r and for small anechoic chambers, the

uncertainties in the power density due to multipath interference and also to

the proximity correction tend to be independent of aperture size. As a result,

the process of "trading off" the uncertainties due to MPI, which tend to

increase with r, and the uncertainties due to C, which tend to decrease with

increasing r, usually leads to values of r roughly between 1 and 1.5. Thus,

all of the variables on the right of eq. (2.8) except A^ are roughly constant

for most situations. Then, the apparent power density obtainable per watt for

a given measurement accuracy is inversely proportional to the physical aperture

area of the antenna.

From eq . (2.8),

fal 2 r2CAP

P = - (KR) . (2.9)

[ b} n For a typical horn antenna, an aperture efficiency of about 0.5 or more can

be assumed; and an a/b ratio of 3/2 or less is a reasonable assumption. If

near-field distances of r = 1 are used, C will usually be about 4/3 or less.

Then for the usual commercial horn antennas,

P < 6 Ap (KR) (2.10)
Comparing the predictions of eq. (2.10) with more detailed calculations for

several horns suggests that eq. (2.10) is fairly conservative. Therefore, it

is believed that eq. (2.11) is justified for likely designs for pyramidal horn

antennas

:

P < 5 A

p (KR) (2.11)

One more consideration is important in arriving at another useful estimation formula. While calibrating EM hazard meters, it is convenient to use

a fixed separation distance for all frequencies within the operating band of

the antenna. If a fixed separation is chosen so that r = 1 for the highest

frequency in the band, r will be about 1.5 for the lowest frequency in the

usual operating band for horns. If the same procedure used to derive eq. (2.11)

is repeated, and estimating that C will be typically less than 1.15 for

separations of r = 1.5, it follows that

PM

< 10 A

p (KR) (2.12)

where is an estimate of the maximum required power at the low end of

(and, therefore, throughout) the operating band for a distance where r = 1 at the high end of the band.

Though eqs . (2.11) and (2.12) are approximate, they are useful for rough

estimates. The major inaccuracy in these formulas is that they over estimate

the required power for horns with small flare angles (because such horns

have much higher aperture efficiencies than 0.5 and also have relatively

small proximity corrections). Also, apertures that have one or more dimensions of about one wavelength or less are apt to have relatively high aperture

efficiencies. For small aperture horns with small flare angles, and particularly for open-ended guides, eqs. (2.11) and (2.12) can over estimate the

required power by a factor of two or more.

The advantages of using a small aperture to minimize the required

oscillator power is particularly pronounced for the case of horn radiators.

If the flare length of a horn is kept fixed while the aperture area is re- duced, the rather large aperture "phase error"* inherent in horns is reduced.

This reduction in phase-error increases the aperture efficiency and also

reduces the proximity correction for a given value of r. Further, if the

flare-angle is reduced to zero, that is for the case of open-ended hollow

waveguide, an unusually large aperture efficiency results that cannot, be

explained by the usual Huygens -Kirchof f antenna theory. For some frequencies

within the operating band, open-ended guides (OEG) have aperture efficiencies

of more than 150 percent (see section 3.1.6).

''•'The "phase-error" is the deviation from uniform-phase illumination. Antennas with uniform-phase illumination are said to be cophasal.

8
2 . 4 Some Conclusions

For most EM hazard meter calibrations it is important to be able to

establish power densities of 10 mW/cm 2 or more, the current maximum permissible exposure level for plane-wave fields. If one wishes to establish

these densities with relatively inexpensive moderate -power oscillators, he

is constrained, at the present time, to approximately the following powers:

60 watts from 100 MHz to 1 GHz, 40 watts from 1 to 1.5 GHz, 25 watts from

1.5 to 2 GHz, and 20 watts from 2 to 20 GHz. Then eq. (2.12) indicates that

many horn antennas are not small enough given the constraints imposed here.

For example, eq. (2.12) indicates that for a 25 watt oscillator Ap should be

no larger than 250 cm2. For frequencies below 2.6 GHz, this is a rather

small aperture, and the availability of such horns is poor or non-existent.

The above discussion, and a perusal of the commercially available horns,

allows some important conclusions. Within the constraints of r > 1 , moderate

oscillator power, and the need to establish power density readings of

10 mW/cm 2

: (1) Most of the popular Naval Research Laboratory [15] design horns

are too large

.

(2) Above 2.6 GHz (above the WR430 band) some horns with sufficiently

small apertures and flare angles are available that have gains of

about 15 dB or more. (The importance of this approximate gain

minimum is explained below.)

(3) From about 2.6 GHz down, available pyramidal horns are increasingly

unsatisfactory due to increasing aperture area.

(4) Because it is highly desirable that the probe being calibrated

be much smaller than the aperture dimensions of the horn,* EM

hazard meters with large probes are relatively difficult to calibrate.

Conclusion (2) above does not follow from eq . (2.12) for the WR284 band (2.6

to 5.95 GHz); however, a more detailed calculation for an available small

flare-angle horn with about 15 dB gain (about 15 by 20 cm aperture dimensions)

shows that 10 mW/cm 2 can be generated with less than 20 watts of power.

At higher frequencies the availability of small aperture horns is better, and

less than 10 watts of power are required for some available horns

.

The importance of having a gain of 15 dB or more is that comparisons

between measured and calculated gain values for such horns has, to date,

always been within ± 0.3 dB . For horns with smaller gain values, our experience at the National Bureau of Standards (unpublished work) has been

that the calculated gain values are likely to be in error by more than

i 0.3 dB . Apertures with dimensions smaller than about one wavelength have

*When using the proximity correction method, it is possible for the field

to be significantly non-uniform over che dimensions of the probe if the probe

is too large compared to the aperture of the radiator.

9
much larger gain values than predicted by the usual Huygen ' s-Kirchof £ theory.

(Open-ended guide has nearly twice the calculated gain at its lower frequency

limit.) Since essentially the same theory is used to calculate near-field

corrections for horns, one would also expect the calculated corrections to

be more accurate for horns with 15 dB or more gain. The accuracy of nearfield calculations is difficult to assess, but it is reasonable to assume uncertainty limits of 10 percent of the decibel value of the correction for

r > 1 [3]. Then for r = 1, nearly all horns will have a corresponding

uncertainty of less than ± 0.2 dB.

Though it would be better to use measured gain and near-zone correction

values for EM hazard meter calibrations, it would appear from the last paragraph that calibration fields with near-field power densities known to well

within ± 1 dB can be established above 2.6 GHz by using calculated values.

It must be emphasized, however, that near-field corrections at points on the

radiation axis of the horn may differ substantially from the proper proximity

correction if the probe of the EM hazard meter is not much smaller than the

aperture of the horn. Most EM hazard probes are quite small, but it is

probably true that not all existing EM hazard meters can be- calibrated under

the stated constraints without some experimentally determined proximity cor- rections. Regardless of these limitations, it is obvious that the existing

calibration capability above 2.6 GHz is much more satisfactory than that

below 2.6 GHz. Therefore, due to the limited funding available, it was decided to concentrate entirely on the frequency region below 2.6 GHz

where satisfactory radiators with easily calculable characteristics are not

avail able

.

2 . 5 Open-Ended Hollow Waveguide Radiators

For frequencies below 2.6 GHz, it is tempting to use open-ended hollow

waveguide (OEG) radiators to establish calibration fields for EM hazard meters.

Some obvious advantages of OEG radiators are:

(1) They are inexpensive.

(2) They can be obtained in long lengths (important for extending the

radiating aperture well in front of an absorbing wall without

having a flange exposed to add to the ever-present scattering

problem)

.

(3) They are rugged, stable, and lightweight.

(4) They are apparently the smallest aperture -type radiators commer- cially available, thus providing maximum field intensity per watt

of source power.

(5) Accurately scaled waveguides are available from the WR430 band to

at least as low as the WR2300 band (0.320 to 0.490 GHz) so that gain

and near-field correction values can be easily scaled.

10
Some other advantages of OEG radiators are:

(6) The gain function of OEG is smooth and nearly linear with frequency.

(7) For most EM hazard meters, particularly those of recent design,

the proximity corrections with respect to OEG will be smooth and

nearly linear with frequency.

(8) Along the radiation axis, at least, the backscatter from fields

incident on OEG is quite small (important for reducing the interaction between the transmitting antenna and probe)

.

Advantage (6) is known from [11] and from the measurements reported in

section 3.1; advantage (7) follows from the fact of smooth and nearly linear

gain and assuming a field sensor small compared to the local field variations

(i.e., small compared to the aperture of the OEG); advantage (8) was determined

as part of the work reported in section 3.1.3.

In addition to the relatively small calibration field volume provided by

OEG, there are two other potential disadvantages associated with the use of

OEG. One of these is the fairly large reflection coefficient of OEG. As

shown in table 1, WR430 has a maximum in-band |r| of about 0.282 (VSWR less

than 1.8) at 1.7 GHz. However, this reflection coefficient magnitude is not

serious since, if not corrected by tuning, existing moderate -power sources can tolerate this degree of mismatch and the forward power loss is less than 8 percent

.

The most serious potential disadvantage of OEG is its relatively low

directivity. The relatively low directivity of OEG causes a strong illumination of the surrounding test chamber which then produces a relatively large

amount of scatter into the test region. However, the resulting multipath

interference level will not necessarily be large at the calibration point be- cause the ratio of the separation between the aperture and the calibration

point to the separation between the aperture and the scattering objects is

relatively small when using small apertures. That is, the calibration point

is isolated from the surrounding chamber by "dispersion" of the radiated

field. Some rough calculations indicate that for the WR430 band the MPI problem

will be less when using OEG in small chambers than when using a 15 dB gain

horn. This assertion is consistent with the measurements reported in

section 3.3.3.

It can be concluded that OEG makes a very satisfactory radiator for

calibrating EM hazard meters with small field sensors when only moderate

source power is available and when small test chambers are to be used.

Some necessary restrictions and techniques when using OEG are discussed in

section 3.3.

11
2 . 6 Summary of Experimental Work Performed

Regarding the discussion in the preceding sections, the experimental

work was limited to that necessary to implement the use of OEG radiators for use in the WR430 band and below. Specifically, the following tasks were performed

:

(1) Gain measurements for open-ended WR430 (and scaling of the results

for WR650, 975 and 1500). (See section 3.1).

(2) Determining near-field corrections for small probes located on-axis

in front of open-ended WR430 (and scaling the results for WR650, 975,

and 1500). (See section 3.2).

(3) A calibration of an EM hazard meter was performed for the WR430 band

to confirm the suitability of the calibration scheme. (See section

3.3).

3. EXPERIMENTAL WORK PERFORMED

3 . 1 Gain Measurements

3.1.1 Introduction

Gain measurements for WR430 guide were performed on a small indoor

range using an extrapolation technique [4]. This technique permits accurate

gain measurements at reduced separations and is often employed by the National

Bureau of Standards to measure gain - standard horns. It is theoretically

rigorous and includes provisions for evaluating both proximity and multipath

interference (MPI) effects. The technique is similar to the usual twoantenna or three -antenna methods [3,4] for absolute gain measurements, but

involves the measurement of the received signal as a function of separation

distance. If the measurement is properly designed [3,4], the recorded data

will clearly show any significant MPI as periodic variations as a function

of separation. These variations are averaged out and the smoothed signalversus-separat ion data is processed in a way that allows an extrapolation

of the signal to "infinite" separation [4]. In the following discussion this

technique is assumed to be familiar, but some further explanation is included

for continuity.

Open-ended pieces of waveguide will be nearly identical in their electrical and radiating characteristics if they are within the usual specified

tolerances (± 0.013 cm for both dimensions of WR430) . Since the aperture

efficiency will be nearly constant for small differences in the aperture

dimensions, the differences in effective receiving area, and thus also the

gain, will be nearly proportional to the differences in physical aperture

area. For WR430 guides within specification, the gains will be within

± 0.015 dB. This small variation justifies the use of the two-antenna method

for the measurements reported here.

12
Insofar as the antennas are identical, their polarizations are identical

so that any polarization mismatch is due merely to rotational misalignment.

Since the polarization patterns are broad and it is easy to accurately align

pieces of OEG, the error due to polarization mismatch can be expected to be

well within 0.01 dB. Further, due to the broad radiation pattern of OEG,

the error due to directional misalignment of the guides can be expected to

be well within 0.01 dB. Then the working formula for the gain measurement

reported here is

G = U

h

N LO

(3.1)

where M is a calculated mismatch factor (see section 3.1.5), d is the separation between the planes of the open ends of the guides, 1 P^ is the power

delivered to the load when the generator and load (see figure 1) are coupled

directly, ^P^q is the power delivered to the load through the antenna transmission path and after any MPI variations are removed (see section 3.1.3), and

N is an experimentally determined correction factor for proximity effects

(see section 3.1.4).

3.1.2 Equipment Used and System Calibration

The pieces of WR430 used for the gain measurements were each about 60 cm long and without flanges on the open-ends. The open ends and the faces of

the flanged ends were machined to provide surfaces precisely orthogonal to

the axis of the guides . The dimensions and angles of the open ends were measured and found to be well within tolerance.

The equipment arrangement is shown in figure 1. This simple measurement

system provided adequate accuracy for measuring the insertion-loss ratio

(^P^/^PjO as needed for the gain determination. The source was a sweeposcillator providing at least 20 mW of power over the WR430 band. Previous

measurements on this source showed no significant spurious frequencies in this

band except for harmonic frequencies, which are removed by the harmonic filter.

The digital frequency meter was checked against a standard frequency and found

to be within one part in 10 6 immediately before the gain measurements. The

digital power meter was used to provide easily known changes in the transmitted power. It was tested and found to be linear within 1 percent over a 20 dB dynamic range by comparing it against an NBS Type II Power Measuring

System [12] (utilizing a single -element bolometer to avoid the error associated

with dual-element bolometers). The "PIN" diode modulator and square-wave driver

were used to provide a modulated signal for the receiving system. The dc

block isolates the rest of the system from any low frequency noise produced by

the PIN diodes and also prevents ground-loop noises when the generator and load

are coupled.

13
The receiving system was designed to be small and light so that it could

be easily moved to the generator for generator - load couplings. The cable between the (tunnel- diode) detector and the 1 kHz amplifier was long enough so that only the hollow-to-coax guide adapter, 20 dB pad, and detector had to

be moved. The logarithmic amplifier was used to provide a more convenient

output signal for recording on the X-Y recorder. The receiving system (the

"load") was calibrated in terms of the X-Y recorder trace by coupling the

generator and load and comparing the receiving system against an NBS Type II Power Measuring System (utilizing a s ingle -element bolometer) that was temporarily substituted for the digital power meter. Over the same 10 dB

range that would be used to record the received signal as a function of

separation, the receiving system was linear to within 0.02 dB . This linearity

test includes any effects due to possible changes in the modulation waveform

as a function of power level

.

The X-axis of the recorder was driven by a voltage proportional to the

separation distance between the apertures of the antennas. This voltage was derived from a mechanically- driven potentiometer connected to the cart supporting the receiving antenna tower (see figure 2) . The X-axis trace of the

recorder was calibrated by using a pointer attached to the cart and measuring

the separation distance by means of a measuring tape placed along one of the

rails. The separation distance could be measured to within the resolution

permitted by the recorder, ± 0.1 percent. Before collecting data to compute

the insertion-loss-ratio versus frequency, the RF power was turned off and

the load was manipulated in the same manner that would be used in these

measurements while observing both the X and Y axes of the recorder. This was done to determine if significant instabilities were present due to changes

in ground loop currents when the generator and load were coupled, 1 kHz noise

leaking into the receiving system, etc. Instabilities from such problems were

well below 0.01 dB.

3.1.3 Insertion Loss Versus Distance

Data were collected to determine the insertion-loss ratio at 11 frequencies; 100 MHz intervals from 1.6 GHz to 2.6 GHz. For each frequency,

with the generator and load connected to the antennas, a "final" transmitted

£

power level was set and the power meter reading was recorded. Then 6 or more recordings of the received signal were made. For each recording, the

absorber blocks behind each antenna were moved to new positions along the

guide, but not advancing the absorbers more than about 15 cm from the rear-most

positions, so that the effects of "forward- scatter" and "backscatter" (see

figure 2) could be observed. Figure 3 shows a typical set of these recordings.

Most antennas have sufficiently small back-lobes in their patterns that

forward-scatter and back-scatter are not important during a gain measurement

(though it is always necessary in a careful measurement to evaluate these effects

14
For OEG , however, these scattering components are quite important, as shown

by the recorded data. Unlike the other important scattering components, the

forward-scatter and back-scatter do not change very much in electrical phase

at the receiving aperture (i.e., the path-length difference with respect to

the direct wave does not change much for these components with changes in

separation), necessitating absorber movements to induce the required phase

changes. In figure 3 the shorter variations, with periods of A/ 2 , in each

recording are due to "inter-antenna" (or "mutual") MPI (see figure 2) and

the offsets between the individual recordings are due to forward -scatter and

back-scatter. Other sources of MPI, such as floor scatter, are apparently

very small. The MPI effects were averaged-out* by drawing curves through

the extremities of the MPI variations and then drawing the locus of points

£ midway between these two curves. This final curve, Sq, can be interpreted

as the signal that would be received in the absence of scattering. Before and

after the group of 6 receiving signal curves were recorded, the generator and

load were coupled, the source power was set to give zero dB on the Y-axis,

and the "initial" power level """P^ of the power meter was noted. (If 1 P^ was significantly different for the before and after readings, the data were

discarded and the insertion-loss measurement was repeated.) Then the ratio

^P^q/^P^ is easily obtained from

-10 log(.

£

? LQ /\) = f

S Q

+ 10 logi\/\) (3.2)

f where P^q has been used to emphasize that the MPI has been averaged-out in

this insertion-loss ratio.

Note that the A/2 variations in figure 3 are no more than ± 0.1 dB even

at the closest separations. This indicates that open-ended guide does not have

a strong scattering component back along its axis for 2.0 GHz. Essentially

the same mutual MPI variations were obtained for the other frequencies within

the WR430 band.

£ For the gain calculations, the value of S n at 24 cm was used. (The

f rest of the Sq curve was used to determine the proximity correction as discuss

later.) The uncertainties in the insertion-loss due to the transmitting and

receiving systems (including drift, connector uncertainties, the resolution

of the recorder, and spurious frequencies) are estimated to be within ± 0.1 dB

.

To this value must be added (a) the uncertainty due to the MPI from the

absorbing walls behind the absorber blocks mounted on the waveguide, and

(b) the uncertainty due to the fact that the "true" extremes of the MPI

from the absorber blocks may not have been recorded. To estimate (a), the

*Strictly, the MPI effects should be averaged-out from a signal proportional

to the received power rather than the log of this signal. However, for

variations of less than a few tenths of a dB , there is negligible error associated with averaging a "logarithmic" recording rather than a "power"

recording

.

15
absorbing walls were moved independently of the absorber blocks and the

perturbations in the received signal were observed to be less than ± 0.05 dB

.

For (b) , it is believed reasonable to assume no more than ± 0.05 dB uncer- tainty at 24 cm. Then the total estimated uncertainty in the insertion-loss

is ± 0.2 dB for any frequency at 24 cm, and the corresponding uncertainty in

the gain will be + 0.1 dB

.

3.1.4 Proximity Corrections

The proximity corrections N (section 3.1) are determined, essentially, by

fitting a polynomial function to the relative received power times distance

squared as the dependent variable and inverse distance as the independent

variable [4] . This function is then normalized by dividing by the leading

(constant) term of the series, and the result is, within the accuracy allowed

by the experimental data, equal to 1/N. Usually, automatic data processing

is used to perform measurements by the extrapolation technique. However, the

necessary equipment was not available for this measurement, and a graphical

analysis was used instead. In this instance, the loss of accuracy involved in

using a graphical analysis is believed to be small.

Figure 4 shows a graphical determination of 1/N for 2.0 GHz. The eight

plotted points were obtained from the relative received power after averaging

out the perturbations due to multipath interference; that is, the power as determined from eq. (3.2). Following the usual practice, the separation

distance is measured in terms of r = d/(a 2 /A). Also note that a horizontal

intercept has been assumed for 1/r = 0. Extrapolation data for (strictly)

cophasal apertures will show a horizontal intercept at 1/r = 0 [13] . Though

OEG radiators are not highly cophasal, due to fairly large edge effects, a horizontal intercept was assumed and allowance was made for this approximation

in the error analysis. Dividing the curve of figure 4 by the intercept of

0.416 at 1/r = 0 gives 1/N for all distances.

The relatively large forward and back-scatter involved complicates the

analysis of the uncertainties for N. Referring to figure 3, the large offsets in the recorded data are due to the forward and back-scatter. However,

these offsets do not have a large effect on N. For example, if the upper and

lower envelopes of figure 3 are used instead of the median curve to determine

N, the value of N for 24 cm separation is changed by only about ± 0.15 dB

.

Therefore, it is estimated that the combined uncertainty due to MPI and the

assumption of a horizontal intercept (the only important sources of uncer- tainty in N) are no more than ± 0.2 dB for 2 GHz and 24 cm separation. However, these uncertainty limits can be reduced by the procedure discussed next.

16
The determinations of the proximity correction N for all the other frequencies were performed as described above. Figure 5 shows the values of

N versus frequency for 24 cm separation. A "least squares" linear function

has been fitted to these values. (Because the gain of OEG is very nearly

linear with frequency, it is reasonable to assume that small proximity correc- tions for OEG will also be nearly linear with frequency.) The values of N given by the linear fit are believed to be within ±0.1 dB , and the cor- responding uncertainty in the gain is ± 0.05 dB . (Linear functions for N versus frequency determined for other distances and scaled for other waveguides

are shown in table 3.)

3.1.5 Mismatch Factor

Rather than using tuners to provide matched conditions for the gain

measurement, measured reflection coefficients were used to calculate the

mismatch factor M in eq. (3.1). If the reflection coefficients of the

transmitting and receiving antennas are identical (not necessarily true

for antennas that have identical radiation characteristics) , then M is given

by [2]

M =

i -r r l 2 li-r r 1 2 Va1 11 At'

i-r nr T | 2 (i-|rJ 2

)

2 (3.3)

V" .G L ' lx_|i Al

where r^, r^, and are (respectively) the reflection coefficients of the

generator, load, and OEG antennas. The use of eq. (3.3) is justified by the

measurements of the reflection coefficients of the pieces of OEG (discussed

below) that showed these coefficients to be very nearly identical.

All of the reflection coefficients were measured with a carefully

calibrated automatic network analyzer which was recalibrated several

times during these measurements. The measurements of r^, and were done

both before and after the insertion loss measurements. For all frequencies,

the before and after measurements were in good agreement (|r| within 0.005).

The measurements for each piece of OEG were repeated six times for

each frequency. During these measurements, the OEG's were radiating into

an anechoic chamber and pieces of absorbing material were mounted around the

guides near the flange end. For the six repeated measurements of the reflection coefficients, the OEG's were moved with respect to the chamber and the

absorbing collars were moved along the guides in increments over about A/2

for the operating frequency. As expected, the measured reflection coefficients

varied because of the changes in the received energy scattered back from the

chamber and the absorbing collar. However, these variations were fairly

small. A variation of ± 0.01 for |r.| and ± 5 degrees for 6 A were the greatest

for any frequency and any scattering configuration. These maximum values

were used for a conservative estimate of the uncertainty limits for the re- flection coefficients. The reflection coefficients for each OEG were

17
indistinguishable, and the values listed in table 1 are the average of the

measurements for both OEG's. The maximum value of | T .

| at 1.7 GHz and the

minimum value at 2.5 GHz were fairly repeatable for both guides and probably

represent the actual variations of |r | with frequency for the OEG's.

The values of M

2 ranged from about 1.05 to 1.1, and the maximum uncerh tamty m M , calculated from the uncertainty limits for the reflection

coefficients, was ± 0.05 dB

.

3.1.6 Results and Discussion

Table 2 gives the results of a least -squares linear regression for the

numeric (not dB) gain versus frequency data for WR430. This table also gives

the corresponding functions scaled for WR650, WR975, and WR1500. The deviation of the data points from the linear function was within ± 0.03 dB , and a quadratic fit to the data was not significantly better. The only significant

uncertainties for each data point are those mentioned at the ends of sections

3.1.2, 3.1.3, and 3.1.4. The simple sum of these estimated uncertainties is

± 0.2 dB (or ± 4.7%). Though the averaging involved in the linear regression

might justify somewhat lower uncertainty limits, the author prefers not to

claim less than ± 0.2 dB for the gains as expressed by the functions of table 2. Gain measurements for OEG are rare. However, some comparisons can be

made to previous measurements. Newell [14] has measured a gain of 6.72 ± 0.2 dB

for open-ended WR650 at 1.4 GHz. The gain formula in table 2 gives a gain of

6.92 dB. The difference between these values may be in part due to the fact

that Newell 's measurement was for an OEG with hollow- to -coaxial guide adapter.

FitzGerrell [11] has measured the gain of open-ended WR90. This waveguide

does not have the 2:1 ratio of inside dimensions characteristic of the wave- guides listed in table 2; therefore, only an approximate comparison can be

made with his results. Using the ratio between the large dimensions of the

guides for scaling, the formulas of table 2 give gain values roughly 0.1 dB

higher than FitzGerrell ' s results.

The gain of OEG is considerably larger than predicted by existing

theoretical calculations. The results of various calculations differ but

usually give values of about 2.0 for WR430 at 1.6 GHz. For example, [5]

yields a value of 1.9, which is much less than the value of 3.71 given by

the formula in table 2. Also interesting is the fact that OEG has an aperture

efficiency greater than 1. The aperture efficiency of WR430 at 1.6 GHz is

1.74, that is the effective receiving area is 1.74 times greater than the

physical aperture area.

3 . 2 Near-Field Corrections

Except in the approximate sense mentioned in the next paragraph, the desired on-axis near- field correction for a radiating antenna cannot be determined

from the proximity correction N for power transfer between two identical

18
antennas. The usual procedure for measuring the near-field correction is to use a small receiving probe moved along the axis of the antenna of interest.

Then the response of the probe is processed in the same manner as for the

measurement of the proximity correction for N during a gain measurement (see

section 3.1.4). Actually, this procedure gives the proximity correction for

power transfer from the antenna to the probe. This proximity correction will

be nearly equal to the near-field correction for the antenna provided (a) that

the near-field configuration does not differ significantly from the far-field

configuration (a reasonable assumption for r > 1) and (b) that the probe has

an antenna or field sensor that is much smaller than the aperture of the an- tenna of interest. For the near-field measurements of WR430 OEG reported here,

a small dipole-diode field sensor was moved along the radiation axis of the

guide. These measurements were difficult to make because of the poor quality

of the existing NBS anechoic chamber. The high MPI level in the NBS chamber

limited the accuracy of the measurements, though the accuracy is satisfactory

witn respect to + 1 dB EM hazard meter calibrations.

As noted in previous work [1], the (decibel) near-field correction

ID Log C for the usual horn antenna at moderately near distances is approximately equal to 5 log N, where N is for the case of the power transfer between

two identical horns. Because the calculation of proximity corrections is based

on essentially the same theory as gain calculations and gain calculations for

OEG are inaccurate, it is by no means certain that this approximation will

hold for OEG. However, within the accuracy of the near-field measurements

performed as part of the work reported here, the approximation seems to hold

for OEG. For example, the near-field correction measurement for WR430 at

20 cm and 2.6 GHz was 0.33 ± .15 dB , which compares fairly well with the value

of 0.41 dB for 5 Log N calculated from the formula of table 3. Measurements

of the near-field correction were made at several other frequencies and these

values also were within about ± 0.1 dB of 5 Log N. Thus, until better near- field correction measurements can be performed for OEG, 5 Log N (where N is

given by table 3) is believed to provide essentially the best values available

for the near-field corrections. The agreement between the measured values for

10 Log C and 5 Log N will justify uncertainty limits of only about ± 40% of the

decibel correction.

3 . 3 Applying the Results

3.3.1 Summary of the Calibration Method

Though the gain and near-field measurements for open-ended guide and other

radiators are difficult, it is easy to use the results for calibrating EM hazard

meters. Employing open-ended guides below 2.6 GHz, a small anechoic chamber

with dimensions about 10 meters long and 5 meters in height and width should

allow calibrations from above 20 GHz down to 500 MHz (if premium quality

19
absorbing material for use down to 500 MHz is used to line the chamber). The

field sensor to be calibrated is placed on-axis in front of the radiator at a near field distance of, roughly, 1 to 2 a

2 /A and the near-field power density

Ls calculated from the net power accepted by the antenna, the gain of 'the antenna,

.mikI the near- field correction:

(See discussion in 2.2 concerning "near-field gain.")

Gain value.', for some commonly available horns are given in [15], nearfield corrections for these horns are given in [1], and near-field gain values

can be easily calculated lor any pyramidal horns using the technique described

in | o | . For open-ended waveguide, tables 4 through 7 in this report give the

mi power ("forward minus reflected") required to establish a 10 mW/cm 2 field

level fur various frequencies at two near-field distances for WR430, WR650,

WRU75, ami WR1500. Note that more than 60 watts of net power are required for

frequencies below 0.9 GHz. for frequencies below 0.9 GHz, the following procedure can be used to calibrate at the L0 mW/cm 2 level: (a) calibrate the meter

at a lower level, say 5 mW/cm 2

, at one of the distances given in tables 6 or

7; (b) move the probe to a closer position on-axis so that a reading of

III mW/cm or a Little more can be obtained; (c) adjust the input power to

achieve precisely the same reading on the instrument that was recorded for the

!> m W/ c in level field and note the net transmitted power; (d) multiply this

net power by 2 (or by 10 divided by the particular mW/cm 2 field level used in

(a J) and increase the net power to this amount; (e) the reading on the meter

should now be the same as would be obtained at the original probe position if

sufficient source power were available to establish a 10 mW/cm 2 level at that

distance. If the probe is too large this procedure may not be valid due to the

difficulties described in section 2.2. However, it is believed there are presently no EM hazard probes that are so large as to invalidate this procedure

for WR975 waveguides and larger. Though definitive information is lacking, it seems very unlikely that a field sensor with largest dimension smaller than

1/4 A would lie too large. In effect, the procedure (a) through (e) extends the near rone calibrat ion method to closer distances where the near-field correction is apt to be substantially different for various probes (unless they are

a 1 .1. very small ) . Some important precautions are given in the next section. With reasonable

care, and given an adequate anechoic chamber, the techniques described here

will allow EM hazard meter calibrations using OEG to within about ± 0.7 dB

total uncertainty limits. This estimate is for the simple sum of the following:

1 0.2 dB for the gain uncertainty, i 0.1 dB for the near-field correction at

the Low end of the frequency band and ± 0.15 dB at the high end of the band;

1 0.2 dB for the measurement of the net power; ± 0.2 dB for multipath interference at the low end of the band and ± 0.15 dB at the high end of the band.

(3.4)

20
These error estimates are about the same as for horn antennas used at 2.6 GHz

in a small anechoic chamber.

3.3.2 Precautions and Suggestions

(.1) The choice of absorbing material for the anechoic chamber is very important.

For a small chamber one wishes to use the minimum necessary absorber

thickness since the absorber thickness can seriously limit the working

volume of the lined chamber. For a 10 m by 5 m by 5 m chamber for use above 500 MHz the best choice is believed to be the solid-pyramidal

foam type with about 61 to 66 cm thickness (24 to 26 inches)

.

(2) The blocks of absorber mounted on the OEG ' s in figure 2 were used to aid

in analyzing the scattering during the OEG gain measurements. They should

not be used during EM hazard meter calibrations since they would neces- sarily be closer to the aperture of the guide than the absorbing wall

behind the aperture and would thus likely increase the forward scatter.

(3) The radiator used for the calibration should extend well in front of the

absorbing wall to minimize the forward scatter. This is particularly

important for OEG radiators. A good practice is to extend the aperture

in front of the wall at least three times the absorber thickness.

(41 The estimate of uncertainties for MPI mentioned at the end of the last

section are based on the observed MPI for isotropic probes placed in

front of OEG at the distances listed in tables 4 through 7. For directive

probes oriented for maximum response, the MPI level will be no greater

than this and should be much less. However, if polarization or response

patterns are recorded for directive probes, fairly large errors could occur for some parts of these patterns. The problems involved in making

accurate polarization or response patterns for EM hazard meters is beyond

the scope of this report.

(5j Often the floor of the chamber will scatter more than the ceiling or walls

because of equipment and walkways located on the floor. Orienting the

L-planes of OEG or horns horizontally will help reduce the floor scatter

by means of the (usually) sharper H-plane patterns of these radiators.

(6) As mentioned in sections 1, 2, it is considered desirable to minimize the

interaction between the sensor and the rest of the meter during EM hazard

meter calibrations. Using horizontal polarization helps to achieve this

condition since it is then relatively easy to orient any instrument cables

orthogonal to the electric vector of the field.

21
(7) Particular!)' when using OEG radiators, the equipment used for the calibration may suffer from interference at the source frequency. The meter being

calibrated may also be so susceptible to interference that a valid calibration cannot be obtained (in which case the meter is useless at that

frequency)

.

(8) Determining the net power by subtracting the measured reverse power from

the measured forward power could possibly cause a significant error due

to scattering back to the radiator from the chamber or from the probe

being calibrated. The energy scattered back to the radiator will, in part,

be received by the radiator and erroneously appear to be part of the waveguide reflected power. For the scattering from a reasonably good anechoic

chamber, the resulting error is small. These errors were evaluated with

respect to a rather poor chamber during the OEG reflection coefficient

measurements (see section 3.1.5) and found to cause changes in the apparent

|rj for the OEG of only ± 0.01. Most EM hazard probes also cause small

scattering. However, good practice requires that a new type of probe that

has not been previously investigated should be moved along the axis of

the radiator to induce perturbations in the reflected power. These perturbations are a direct measure of the errors resulting from the scattering

from the probe (unless they are averaged out as a part of the calibration

procedure) .

3.3.3 Test Calibrations

A small isotropic probe was available for the following tests. This probe

was calibrated at 2.5 GHz both with WR430 OEG (at 20 cm) and with a "15 dB"

gain WR284 horn whose gain was known from previous measurements. The two

calibrations agreed to within 0.2 dB, well within the ± 0.5 dB uncertainty

estimate for the calibration using the horn. These measurements were made in

a chamber with dimensions of only about 5 m by 2 . 5 m by 2 . 5 m

.

During the calibration with WR430 OEG, the MPI was measured by moving the

probe along the OEG axis and by moving the absorbing wall behind the OEG.

The total MPI was less than ± 0.2 dB throughout the band. For comparison,

a "15 dB" gain WR430 horn was substituted for the OEG and the MP I was found to

be somewhat larger, ± 0.25 dB, for an equivalent separation in terms of a

2 /A.

(These measurements tend to confirm the discussion at the end of section 2.5.)

For the Lower frequency horns, there will likely be considerably more advantage in using OEG rather than herns.

22
4. REFERENCES

[I] Bowman, Ronald R. and William E. Jessen, Calibration techniques for RAMCOR

densiometer antennas, NBS report (unpublished),

[2] Bowman, R.R., Near-zone calibrations for antennas and field intensity

meters, Rome Air Development Center Technical Report No. RADC-TR- 69-415

(January 1970)

.

[3] Bowman, R.R., Field strength above 1 GHz: Measurement procedures for

standard antennas, Proc . IEEE, 5_5, 981-990 (June 1967).

[4] Newell, Allen C, Ramon C. Baird and Paul F. Wacker, Accurate measurement

of antenna gain and polarization at reduced distances by an extrapolation

technique, IEEE Trans, on Antennas and Propagation, AP- 21 , No. 4, 418-431

(July 1973)

.

[5] Chu, L.J., Calculation of the radiation properties of hollow pipes and

horns, Journal of Applied Physics, 11_, 603- 610 (Sept. 1940).

[6] Jull, E.V., Finite-range gain of sectoral and pyramidal horns, Electronics Letters, 6, No. 21, 680-681 (October 15 , 1970).

[7J IEEE Test Procedures for Antennas, No. 149 (Revision of IRE 2S2) (January

1965), IEEE Trans, on Antennas and Propagation, AP-13 , 437-466 (May 1965).

[8] Crawford, M.L., Generation of standard EM fields for calibration of power

density meters 20 kHz to 1000 MHz, NBS I R 75-804 (January 1975).

[9] Baird, R.C., Methods of calibrating microwave hazard meters, Biologic

Effects and Health Hazards of Microwave Radiation , Proc. of an Intl.

Symposium Warsaw, 228-236 (October 1973).

[10] Soejima, T. , Fresnel gain of aperture aerials, Proc. IEE (London), 110,

1021 -1027 (June 1963) . [II] FitzGerrell, R.G., Swept - frequency antenna gain measurements, IEEE Trans,

on Antennas and Propagation, AP-14 , 173-178 (March 1966).

[12] Larsen, N.T. and F.R. Clague, The NBS Type II Power Measurement System,

1970 ISA Annual Conf. Proc, Philadelphia, PA, Paper No. 712-70,

25, Pt. 3, 12 pages (Oct. 1970).

[13] Wacker, P.F., Electromagnetics Division, National Bureau of Standards,

Boulder, Colorado, Private Communication.

[14] Newell, A.C., Electromagnetics Division, National Bureau of Standards,

Boulder, Colorado, Private Communication.

[15] Slayton, W.T., Design and calibration of microwave antenna gain standards Naval Research Lab., Washington, D.C., Rept. 4433 (Nov. 1954).

23
Table 1. Measured reflection coefficients of WR430 openended guide, 58.8 ± 0.1 cm long.

Frequency |r^| 6^

(GHz) (Dimensionless) (Degrees)

(±0.0001) (±0.01) (±5)

1.6 . 262 -165 .0

1.7 . 282 -56 .6

1.8 .279 73.4

1.9 . 265 -136 .5

2.0 .266 23.8

2.1 .245 -167.9

2.2 .248 8.2

2.3 .232 -172.6

2.4 . 223 11. 6

2.5 .207 -156.9

2.6 .214 31 .1

Table 2. Gain formulas for various waveguides.

Waveguide Gain Formula

WR43 0 G 430

= -0 .054 + 2 353 F

WR65 0 G 650

= -0 .054 + 3 557 F

WR9 75 G

9 75

= -0 .054 + 5 .335 F

WR1500 G 1500

= -0 054 + 8 .208 F

Notes : The formulas are for numeric gain (not dB) . The

frequency F is in units of GHz

.

24
Table 3. Proximity corrections for various waveguides.

d(cm) Formula d [cm) Formula

20 N = 1.0072 + 0.0764 F 30 N = 1.0072 + 0.1155 F

24 N = 1.0028 + 0.0499 F 36 N = 1.0028 + 0.0754 F

28 N = 0.9896 + 0.0392 F 42 N = 0.9896 + 0.0593 F

32 N = 0.9967 + 0.0256 F 48 N = 0.9967 + 0.0387 F

WR 97 5 WR 150 0

d(cm) Formula d(cm) Formula

45 N = 1.0072+ 0.173 2 F 70 N = 1.0072 + 0.2665 F

54 N = 1.0028 + 0.1131 F 84 N = 1.0028 + 0.1741 F

63 N = 0.9896 + 0.0889 F 98 N = 0.9896 + 0.1367 F

72 N = 0.9967 + 0.0580 F 112 N = 0.9967 + 0.0893 F

25
Table k. Net power P required to establish 10 mW/em at two distances on-axis

in front of open-ended WR 1+30 waveguide. Also shown are the (numeric)

gain G, near-field reduction l/C, and the near-field gain G/C.

R = 20 CM

Frequency

(GHz)

1.60

1.70

1.80

1.90

2.00

2.10

2.20

2.30

2.1+0

2.50

2.60

R = 2k CM

Frequency

(GHz)

1.60

1.70

1.80

1.90

2.00

2.10

2.20

2.30

2.1+0

2. 50

2.60

P G l/C G/C

10

(Watts)

ik.ko

13.58

12.86

12.22

11.61+

11.11

10.6U

10.20

9.81

9.1+1+ 9.10

3.71

3.95

1+.18

1+.1+2 1+.65

1+.89

5.12

5.36

5.59

5.83

6.06

0.9^

0.9k

0.93

0.93

0.93

0.93

0.92

0.92

0.92

0.91

0.91

3.1*9

3.70

3.91

l+.ll

1+.32

1+.52

1+.73

h.93

5.13

5.32

5.52

P

10

G l/C G/C

(Watts)

20,,30 3.• 71 O.96 3.57

19..13 3.• 95 O.96 3.78

18,• 09 1+.,18 0.96 1+.00

17.• 17 1+.,1+2 0.95 1+.22

16,.31+ 1+.• 65 0.95 1+.1+3 15.• 59 1+,.89 0.95 k.6k

lit.• 90 5.,12 0.95 1+.86

ll+,.28 5.,36 0.95 5.07

13. 71 5.• 59 OM 5.28

13..19 5..83 0.9k 5.^9

12. 70 6,,06 0.9!+ 5.70

26
Table 5- Net power P^ required to establish 10 mW/cm at two distances on-axis

in front of open-ended WR 650 waveguide. Also shown are the (numeric)

gain G, near-field reduction l/C, and the near-field gain G/C.

R = 30 CM

Frequency

(GHz)

p

io

(Watts)

G l/C G/C

1.00

1. 10

1.20

3U.21

31.21

28.72

3.50

3. 86

k..21

0.9k

0.9U

0.93

3.31

3.62

3. 91+ 1.30

1.1+0

1.50

26.62

2U.82

23.26

^•57

k.93

5.28

0.93

0.92

0.92

1+.25

1+.56

1+.86

1.60

1.T0

1.80

21.90

20.70

19.6k

5.6U

5-99

6.35

0.92

0.91

0.91

5.16

5.1+6

5.76

R = 36 CM

Frequency P G l/C G/C

(GHz) (Watts

1.00 1+8.28 3.50 0.96 3.37

1.10 1+3.98 3.86 0.96 3.70

1.20 1+0.1+1 k.21 O.96 1+.03

1.30 37-39 U-.57 0.95 ^.36

1.1+0 3I+.8I 1+.93 0.95 U . 68

1.50 32.58 5-28 0.95 5-00

1.60 30.62 5. 61+ 0.9k 5-32

1.70 28.90 5.99 0.9^ 5.63

1.80 27.37 6.35 0.9h 5-95

27
Table 6. Net power required to establish lOmW/cm at two distances on-axis

in front of open-ended WR 975 waveguide. Also shown are the (numeric)

gain G, near-field reduction l/C, and the near-field gain G/C.

R = 1+5 CM

Frequency

(GHz)

0.70

0.75

0.80

0.85

0.90

0.95

1.00

1.05

1.10

1.15

1.20

R = 5h CM

Frequency

(GHz)

0.70

0.75

0.80

0.85

0.90

0.95

1.00

1.05

1.10

1.15

1.20

P

10

G

(Watts)

73.>+5 3.68

68.74 3.95

6h.6k 4.21

61.02 4.48

57.81 4.75

54.93 5.01

52.35 5.28

50.02 5.55

47.90 5.81

1+5.96 6.08

Uh. 19 6.35

l/C G/C

0.94 3.46

0.94 3.70

0.93 3.94

0.93 4.17

0.93 4.4o

0.92 4.63

0.92 4.86

O.92 5.09

0.91 5.31

0.91 5-54

0.91 5.76

P

10

(Watts)

103.56 3.68

96.81 3-95

90.92 4.21

85.73 4.48

81.12 U . 75

77.00 5-01

73.30 5.28

69.95 5.55

66.91 5-81

6)4.13 6.08

61.59 6.35

l/C G/C

O.96 3-54

O.96 3.78

0.96 U .03

0.95 4.27

0.95 4.52

0.95 4.76

0.95 5.00

0.94 5-24

0.94 5.^8

0.94 5.71

0.94 5-95

28
2

?able 7- Net power P required to establish 10 mW/cm at two distances on-axis

in front of open-ended WR 1500 waveguide. Also shown are the (numeric)

gain G, near-field reduction l/C, and the near-field gain G/C.

I = 70 CM

Frequency

(GHz)

P

10

(Watts

)

G i/c G/C

o Us

0.50

0.55

1 7Q 6l

162.36

1U8.28

U.05

U.U6

0 Qii

0.9^

0.93

3.79

U.15

0.60

0.65

0.70

136.57

126.68

118.20

k.Q-J

5.28

5.69

0.93

0.92

0.92

U.86

5-21

0.75 11: . £" 6.10 0.91 5.55

Frequency

(GHz)

p"10

(Watts

)

G i/c G/C

0.U5

0.50

0.55

253.31

228.56

208.36

3.6U

4.05

h.h6

0.96

0.96

0.95

3.50

3.88

4.26

0.60

O.65

0.70

191.55

177.36

165.21

I+.87

5.28

5.69

0.95

0.95

0.9^

U.63

5.00

5-37

0.75 15^.70 6.10 0.9^ 5.73

29
1

DIGITAL POWER

METER

IkHz

_n_n_

COUPLER MODULATOR DC

BLOCK

Generator

r

RECEIVER TRANSMITTER

20

dB

PAD

WR 430 Guides

~1

DETECTOR

IkHz

AMPLIFIER

Load

RANGE

POTENTIOMETER

20

dB

PAD

1

L

LOG

AMPLIFIER

X-Y

RECORDER

lgure 1. Block diagram of equipment used for the measurement

of the gain of open-ended WR430 waveguide.

30
o

If) &

CD 4-* U -H U 3 3O T3

[/) CD

+J <D C >

OS

O

CJ

!hO

O

P. </> e r-l •H ni

CD CD x t-. +J 0)3

00

Pi

M

•H

C 3 -H O rH X -H

71 CD(J »—1

PiO *d • H

•H

C t-1 *-> 03 CD

n3 P

S-i in cS 3 i-H e

•i-) nj bO m 3 C C •H O CD J=) (J ^ Sh H O

CD </> M ja S • nj

nj bO M C

•H o

n! Sh •H d <d +J C 4-> ca

CD 4-> •H

fi

a nj < (/>

(SI

<D

<m3

QO •H

tn

31
(gp) jaMOd paAiaoay aAjpiay

32
Figure 5. N versus frequency for 24 cm separation.

33
NBS-1 14A (REV. 7-73)

U.S. DEPT. OF COMM.

BIBLIOGRAPHIC DATA

SHEET

1. PUBLICATION OR REPORT NO.

NBSIR 75-805

2. Gov't Accession

No.

3. Recipient's Accession No.

4. TITLE AND SUBTITLE

Calibration Techniques for Electromagnetic Hazard Meters;

500 MHz to 20 GHz

5. Publication Date April 1976

6. Performing Organization Code

276.05

7. AUTHOR(S)

Ronald R. Bowman

8. Performing Organ. Report No.

9. PERFORMING ORGANIZATION NAME AND ADDRESS

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

WASHINGTON, D.C. 20234

10. Project/Task/Work Project/ l a Unit No. 2765279

11. Contract/Grant No.

12. Sponsoring Organization Name and Complete Address (Street, City, State, ZIP)

Calibration Coordination Group

Army /Wavy /Air Force

13. Type of Report & Period Covered

14. Sponsoring Agency Code

15. SUPPLEMENTARY NOTES

16. ABSTRACT (A 200-word or less factual summary of most significant information. If document includes a significant

bibliography or literature survey, mention it here.)

The calibration techniques discussed are suitable for producing fields for calibratinj

most electromagnetic (EM) hazard meters to within ± 1.0 dB using a minimum of laboratory]

space and oscillator power. Above about 2.6 GHz, adequate equipment and standards have

been available for these calibrations. Below this frequency the large apertures of the j usual horn radiators require more power than is available from medium power oscillators.

Further, calibrations in closed systems are difficult except at frequencies well below

1 GHz. Thus there is a need for small-aperture gain standards from about 500 MHz to

2.6 GHz. The main portion of the work reported here consists of accurate gain measurements for open-ended hollow waveguide radiators (OEG) for use from 500 MHz to 2.6 GHz.

Other characteristics of this type of radiator important for EM hazard meter calibration

were also determined: near-field corrections, reflection coefficients, and aperture

scattering. The suitability of the calibration scheme was tested by performing calibrations at 2 GHz on an EM hazard meter with both a horn radiator and an OEG radiator.

17. KEY WORDS (six to twelve entries; alphabetical order; capitalize only the first letter of the first key word unless a proper name; separated by semicolons

Calibrations; electromagnetic hazards; field meters; gain; microwave; near-field.

18. AVAILABILITY \K' Unlimited

^ For Official Distribution. Do Not Release to NTIS

"~

' Order From Sup. of Doc, U.S. Government Printing Office Washineton. D.C. 20402. SD Cat. No. C13

jX

! Order From National Technical Information Service (NTIS)

Springfield, Virginia 22151

19. SECURITY CLASS

(THIS REPORT)

UNCLASSIFIED

21. NO. OF PAGES

37

20. SECURITY CLASS

(THIS PAGE)

UNCLASSIFIED

22. Price

$3.75

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