NBSIR 82-1663

IMPROVED COAL INTERFACE

DETECTOR

Keith C. Roe

Ronald C. Wittmann

National Bureau of Standards

U.S. Department of Commerce

Boulder, Colorado 80303

May 1982

NATIONAL BUREAU OF STANDARDS

LIBRARY FEB 2 1983

/?csrCLCC, - NBSIR 82-1663

IMPROVED COAL INTERFACE

DETECTOR

Keith C. Roe

Ronald C. Wittmann

Electromagnetic Fields Division

National Engineering Laboratory

National Bureau of Standards

U.S. Department of Commerce

Boulder, Colorado 80303

May 1982

Final Report

Prepared for: Department of Energy

Pittsburgh Mining Technology Center

Pittsburgh, Pennsylvania

U.S. DEPARTMENT OF COMMERCE, Malcolm Baldrige, Secretary

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director
'

. .

. • v J-.
TABLE OF CONTENTS

Page

Abstract 1

Executive Summary 1

I. Background 3

II. Operation of the FM/CW System 4

III. Theory 5 111.1 The System Response Function r(t) 7 111. 2 r(t) and the Reflection Coefficient, R(w) 8

111. 3 Response From a Dielectric Interface 10

111. 4 Closed Loop System 13

111. 5 Resolution 13

111. 6 Tuning the FM/CW System 14

111. 7 Numerical Examples 16

IV. Instrumentation 17

IV. 1 Microwave System 17

IV. 2 Signal Processing System 18

V. Test Facility 19

VI. Target Material 21

VII. Testing 21

VI 1. 1 Tracking a Changing Dielectric 22

VII. 2 Measuring Through Coal Thicknesses Up To 38 cm (15 in) 22

VII. 3 Tracking the Interface Independent of Height 25

VI 1. 4 Acquiring and Tracking the Coal -Shale Interface in the

Environment of Metal Surfaces 26

VIII. MSHA Approval and In-Mine Tests 26

IX. Project Conclusion 27

Acknowledgments 27

X. References 28

i i i
LIST OF TABLES

Page

Table I. Dielectric Constant and Loss Tangent of Simulated Coal Target

Material in the Frequency Range 1 to 4 GHz 23

Table II. Thickness Measurement Results Using the CID System and

Simulated Coal Target Material 24

iv
LIST OF FIGURES

Page

Figure 1. A simple FM/CW radar block diagram. "D" denotes a divider, "M" denotes a mixer 29

Figure 2a. A triangular frequency versus time relationship useful in FM/CW

radar applications 30

Figure 2b. The reference signal and the target signal are displaced in

time by an interval, t, which is the difference in travel

times for the two paths of figure 1. The displacement in time

corresponds to a displacement of frequency, The FM/CW

system detects this "beat" frequency 30

Figure 3. Use of a phase discriminator for the detection of the real

and imaginary parts of r(t) 31

Figure 4a. Reflection from an infinite plane boundary between two media.... 32

Figure 4b. A primary ray path (solid line) and a secondary ray path

(dotted line). Secondary paths result from multiple reflections

within the medium 32

Figure 5. This figure shows a hypothetical response, r(t), from a planar

interface. r(t) is basically sinusoidal, but the slope is

discontinuous where the frequency ramp changes slope. These

discontinuities contribute sidebands to the spectrum of r(t) 33

Figure 6. Simulation of an FM/CW system responding to an interface at

various distances. The sweep range is from 1-2 GHz. Shown

is the spectral amplitude at each of the first 30 harmonics 34

Figure 6. Continued 35

Figure 7. Simulation of an FM/CW system responding to an interface at

various distances. The sweep range is from 1-3 GHz. Shown

is the spectral amplitude at each of the first 30 harmonics 36

Figure 7. Continued 37

Figure 8. Block diagram of the FM/CW system used as a coal interface

detector 38

Figure 9. Schematic circuit of the triangular wave generator used in

the FM/CW system 39

Figure 10. A copy of the computer results and display from a CID Measurement 40

Figure 11. Photograph of the CID test bed facility 41

v
Page

Figure 12. Effect of antenna to interface distance on thickness

measurements 42

Figure 13. Results of measurements made in the presence of metallic

surfaces. Solid line represents the top surface of the

dielectric material 43

Figure 14. Block diagram of the FM/CW system proposed for an experimental

permit. The power cable and fiber optic cable (50 meters long)

go to the spectrum analyzer and power source located in a non-hazardous area of the mine 44

vi
IMPROVED COAL INTERFACE DETECTOR

This report describes the theory, design, construction and

testing of an electromagnetic coal interface detector. The

purpose of this type detector is measuring the thickness of roof

coal left during underground mining operations. An above ground

test facility constructed to evaluate the coal interface detector

is also described.

Key words: coal-shale interface; electromagnetic coal interface

detector; FM/CW radar; roof-coal thickness.

Executive Summary

In many underground coal mines a specified thickness of coal is required

to be left in the roof as the coal is mined to maintain roof stability and

prevent exposure of the overburden to air that might cause it to

deteriorate. Determining the thickness of the coal left in the roof during

mining is important for safety reasons and to maximize profit to the mine

operators.

The need for roof coal measurement systems becomes important as automated

mining machines come closer to reality. Historically, the roof-coal thickness

has been determined by drilling holes through the coal or by observation of

coal strata or partings that in some mines are at known distances from the top

coal -shale interface. In recent years other systems have or are being

developed to monitor this thickness. (One other system measures natural

radiation from the overburden that is attenuated as it passes through the

coal.) The system (FM/CW radar) described in this report calculates the coal

thickness from time delay measurements of electromagnetic waves reflected from

the bottom and top surfaces of the roof coal.

In an FM/CW radar system the rf or microwave source is frequency

modulated, producing a continuous wave signal whose frequency changes linearly

with time. The signal is split so part of it travels through a fixed path

length and then is mixed with the other part that has been transmitted and

reflected from the interface. Since the reflected signal travels a different

path (usually longer) and the frequency source is frequency modulated the

mixed signal contains a difference frequency that is proportional to the rate

of modulation and the difference between the two path lengths. Signals

reflected from one or more reflecting surfaces produce one or more separate

frequency differences which are proportional to the propagation path lengths

1
to each reflecting surface. Detection and identification of those difference

frequencies yields the information necessary to determine the distance to each

reflecting surface and therefore, the distance between the two reflecting

surfaces needed in a roof-coal thickness measurement.

The work described in this report follows an earlier effort to produce a coal interface detector (CID) system. In that first system the frequency

differences were detected in a bank of fixed bandpass filters. The readout

was a digital counter set to turn on and off from frequency responses coming

through the filter bank. (For details refer to DOE report FE/8881-1). That

system was designed to be as simple as possible so that an untrained operator

could use it. However, a reading took anywhere from 15 to 30 seconds and

continual adjustments were required as the coal thickness changed or the

distance from the antennas to the air-coal interface changed. The net result

was that the readings came too slow to be used on a continually moving mining

machine and a full time operator was required in order to use it.

This project was set up to improve the earlier CID system by providing a system that would respond faster (2 or 3 seconds) and with some built-in

decision making capability (a computer) to automatically choose the frequency

responses corresponding to the top and bottom surfaces of the roof coal.

In the first CID project testing was done in underground coal mines.

Those sites were in areas assigned by the coal mine operators where the coal

had been mined many years (up to 30) previously. The roof coal was sagging or fallen away and did not represent roof coal in a freshly mined area. The coal

thickness was determined by drilling through the coal at selected locations

and comparing the measured thickness with the readings obtained from the CID

system. Those tests were very limited in the number of measurement points and

in the parameters of the coal being measured. No definite information was available about the dielectric constant and the attenuation of the coal.

The testing capability was expanded in this project to provide an above

ground test facility for the CID system using simulated coal targets with

known thicknesses, attenuation, and dielectric constant.

This second CID project was organized in two phases. The first phase was

to develop the improved CID system and perform the above ground testing using

the test facility and simulated coal target material. The second phase called

2
for getting an experimental permit from the Mine Safety and Health

Administration (MSHA) and performing in-mine tests with the CID system at or near the working areas of coal mines where the coal has been recently cut.

The following sections of this report describe in more detail the CID

system, the theory, the above ground test facility, and the testing efforts.

I. Background

Roof-coal thickness measurements by Ellerbruch, Adams, and Belsher [1-2]

revealed variations of coal relative dielectric constant ranging from four to

sixteen. These variations were probably caused by moisture variation from

water seepages that were present in the measurement area in some areas but not

in others.

Their measurements also showed that the rf attenuation through coal was approximately 40 dB/m (1 dB per inch) at 1 GHz. The attenuation doubles for

each octave increase in frequency so that at 4 GHz the expected attenuation of

microwave signals through the coal is 160 dB/m.

Those experiments formed the basis for most of the work done in this

project and the earlier projects [1,2] to develop an electromagnetic coal

interface detector (CID).

Several electromagnetic techniques were investigated by Ellerbruch and

Belsher [2]. The most promising were a pulsed radar system and an FM/CW

system. A limitation to the pulsed system was a lack of antennas that were broadband enough to handle the pulsed signals efficiently and that had enough

directivity (focusing capability) to be useful. Resolution of the pulsed

radar was also a problem since a resolution of 2 or 3 centimeters (» 1 inch)

requires a pulse width of about 100 picoseconds. Pulse generators capable of

delivering 100 picosecond pulses with peak voltages of 25 to 50 volts were not

available. The FM/CW system was chosen over the pulsed system because

components necessary for such a system were available. The expected

resolution of an FM/CW system operating from 1 to 4 GHz was approximately 2 to

3 centimeters. (Factors limiting resolution are described in section 1 1 1. 5.)

Since the earlier measurements showed a possible variation of dielectric

constant in the coal, a requirement was that the CID system automatically

3
correct the coal thickness measurement for any changes in the coal dielectric

constant.

Testing and demonstrating the capabilities of a coal interface detector

was a major task in this program. Because of the need for extensive testing

during developmental stages of the CID and the problems of locating and

getting in and out of underground test sites, an above ground test facility

was constructed. This facility was used during assembly of the CID system and

for verification tests afterward.

II. Operation of the FM/CW System

A simplified diagram of an FM/CW system is shown in figure 1. Power from

a frequency modulated microwave generator is divided into a target path and a reference path. The reference path is brought directly to the mixer while the

target path is routed to the target and back via a transmitting and a receiving antenna. The received signal is fed into the remaining mixer

input. The spectrum of the mixer output is displayed on a spectrum analyzer.

Assume that the frequency versus time character of the signal is as shown

in figure 2a. At the mixer, the received signal and the reference signal will

be displaced in time by an interval, t, which is the difference in transit

times for the target and reference paths (see figure 2b). The displacement in

time is equivalent to a displacement in frequency, u^,

= ax (1)

where a is the rate of change of frequency in time. (Turning points in the

ramp have been ignored for the present.) The difference frequency, is the

low frequency component of the mixer output. For a general target, of course,

there will be many difference frequencies. Each one will correspond to the

delay incurred by the incident wave as it is scattered from the various parts

of the target. If the major scattering regions are sharply defined and wellseparated, the difference frequencies will show up as a series of peaks in the

spectrum of the mixer output. The fact that and x are related by (1)

allows us to calibrate the FM/CW system to indicate the ranges of these

scatterers.

4
Typical values of the parameters for systems we will be discussing are:

2aw/2tt = 1 GHz (bandwidth)

P = 0.1 - 0.01 s (sweep period)

t = 1 - 100 ns (delay)

a = 10 - 100 GHz/s.

Thus, the difference frequency, will be on the order of several kHz or less.

The heuristic discussion of FM/CW operation, given above, will be made more precise in the following paragraphs. In the first place, since the

bandwidth is limited, the spectrum of the mixer output will not in general

consist of a single frequency, u^, even when only one delay is involved.

Rather, the spectrum will be spread out around This broadening, which

depends solely on the bandwidth, leads to an expression for the ultimate

theoretical resolution of an FM/CW radar. Another point worth noting is that,

since the system is driven periodically, the observed spectrum is discrete

with contributions only at harmonics of the sweep rate.

The choice of the triangular wave time-versus-frequency relationship, as

in figures 2, is worth noting. Without linearity the concept of and its

simple relationship to the delay, t, loses meaning. One could also use a sawtooth waveform, where the negative slope portion of the ramp is reduced to

negligible duration and blanked, but the discontinuities introduced at the

transitions between the top and the bottom of the band are very difficult to

track with real hardware, and the ringing that necessarily results can degrade

system performance. The triangular wave appears to be the best compromise for

a linear time versus frequency device operating over a finite bandwith.

III. Theory

In order to proceed we will need to state precisely what will be meant by

the FM/CW wave. To do this it is necessary to define amplitude, frequency,

and phase for signals which are not pure sinusoids.

Consider the general, real signal, s(t). The corresponding analytic

signal is defined as [3]

5
S(t) = s(t) + i s(t)

.

s(t) is the Hilbert Transform of s(t), or, in other words, s(t) is the signal

resulting when s(t) is passed through an ideal 90° phase shifter. The

magnitude of the analytic signal, |S(t)|, is called the "envelope function"

and may be regarded as a suitable generalization for the amplitude of a nonsi nusoi dal function. The concept of an envelope function is most meaningful

if | S ( t ) I varies much more slowly in time than s(t). In this case the power

in s(t), averaged over a time long compared to variations in the signal and

short compared to variations in the envelope, is proportional to I S ( t )

1

2 [3],

It is most convenient to consider signals which have amplitudes of unity

(i.e., signals that have unit envelope functions). Such functions may be

represented quite generally as

S(t) = expi$(t).

$>(t), which will be referred to as the phase function, cannot be chosen

arbitrarily as it is also a property of an analytic signal where its spectrum

vanishes for negative frequencies. This condition may be stated as follows:

expi$>(t) c(w)expiwt dw. ( 2 )

where the integral runs only over positive w. An example of such a signal is

a pure sinusoid, $>(t) = w

o

t, whose spectrum contains only one frequency, w

Q

.

The instantaneous frequency, w(t), which will usually be referred to as simply "frequency," is defined to be

u)(t) = (3)

While it is not within the scope of this report to discuss in detail what

constraints must be imposed on $(t) in order that eq (2) be satisfied [4], it

may be stated quite generally that if the frequency u>(t) varies little over

the oscillation period 2ir/w(t), then eq (2) will be wel 1 -approximated.

6
Mathematically this may be stated as

a/w2 <<l

.

(Remember: u> = d$/dt, a = d

2 $/dt2

). Physically, this condition means simply

that, locally in time, the signal may be regarded as a pure sinusoid of

frequency, w(t).

10 9 _ 8 For typical system parameters (a ~ 10 Hz/sec, u ~ 10 Hz), a/w2 ~ 10 ; thus, any errors resulting from approximating the analytic signal

by expi$(t) are expected to be negligible.

To summarize, for the purposes of this report an FM/CW wave will be

assumed to have the form:

expi«(t), (4)

where $(t) is subject to the condition

a/w2 << 1. (5)

We will take the real part of the analytical signal eq (4), cos$(t), as our primary signal. The imaginary part of eq (4), sin$(t), which is cos$(t)

shifted by 90° in phase, will also be of interest in the following discussion.

III.l The System Response Function, r(t)

Referring again to figure 1, let us assume that the signal in the

reference path is cos$(t). The wave returning through the signal path has

been modified by the target and may be written, in all generality, as:

Re[r( t)expi$(t)] = A(t)cos$(t) - B(t)sin$(t) (6)

where r(t) = A ( t) + i B(t). Note that r(t) contains all of the available

information concerning the nature of the target.

The mixer output is proportional to the product

7
cos$(t)[A( t)cos$(t) - B(t)sin$>(t)]

=1/2 A(t) + 1/2 A(t)cos2$>(t) - 1/2 B(t)sin2*(t)

.

The high frequency component of the above equation may be filtered out and

ignored, leaving only 1/2 A(t) as the output. (The fact that A(t) is "low

frequency" may be guessed from the discussion of II where it was suggested

that the spectrum of the mixer output is important only near the difference

frequencies. The "low frequency" nature of A(t) will be made clearer by

specific examples in the following sections.)

If the reference signal were passed through a 90° phase shifter before

being applied to the mixer, then the output would be the low frequency

component of the product

s i n ( t ) [ A ( t)cos<i>(t) - B(t)sin$(t)]

or,

= -1/2 B(t) + 1/2 B(t)cos2*(t) + 1/2 A(t)sin2*(t)

,

-1/2 B(t).

The function, r(t), will be referred to as the ( FM/CW ) response

function. A detection scheme for measuring both A(t) and B ( t ) simultaneously

is shown in figure 3, where Q denotes a quadrature divider (one output is

shifted 90° in phase). The output proportional to A(t) will be called the

cosine port output, while the output proportional to B(t) will be called the

sine port output.

III. 2 r(t) and the Reflection Coefficient, R(w)

Under this heading we will assume that the various waves involved in the

analysis, including the fields radiated by the antennas and reflected by the

target, are TEM in nature. With this condition, at a given frequency, the

fields scattered back to the receiving antenna may be specified in terms of

the incident wave and the plane wave reflection coefficient, R(u>). It is the

goal of the following paragraphs to show how the response function, r(t), is

8
related to R(w). The assumption of TEM waves for the radiating spectrum is

not expected to affect the qualitative nature of the result.

Let us consider the transmitted signal written in the form

oo expi$(t) = / expi$(t') 6 ( t- 1

' ) dt 1

-oo

where 6 ( t) is the Dirac delta function. 6 ( t ) is the ideal impulse function,

being "equal" to zero everywhere except at t = 0 where it is unbounded; yet

the singularity is such that <5(t) remains integrable, and:

e/

6 (

t

1 ) dt ' = 1, -e

where e is an arbitrarily small positive number. These properties serve to

establish the above representation of expi$(t). One way of writing <5(t),

which is a consequence of Fourier Theory, is [4]

1 6 ( t) = I expiaj'tdo) 1 . —00

This represents a decomposition of S(t) into plane waves. (Note the + iwt

time dependence.)

The reflected signal may now be written as

OO oo r(t)expi$(t) = / dt'expi$(t') x J R(w') expiV ( t-t' ) dco

1

= / expi$(

t

1 )h( t-t' ) dt 1

—oo

00 = / expi$(t-t' )h( t' ) dt '

,

o

where h(t) is the impulse response function and

1 h(t)=

2

^- /

R(oj') expiw't dw 1 . (7)

—oo

The impulse response, h(t), will have significant amplitude for a limited

period of time after the occurence of the exciting impulse. This time period

may usually be taken to be the delay correspond!" ng to the most distant

9
scatterer, although it might be necessary to consider multiple reflections

among scatterers and dispersive effects, which will increase the duration of

the response. If the response duration is short enough, we may write

oo r(t) = / exp-i[>( t)-$( t- 1

' ) ]h( t' )dt ( 8 )

oo « / exp-iu)(t)t‘ h (

t

1 ) dt '

,

where we have used the Taylor Series approximation

$ ( t-t

1 ) » $( t) - o)( t)t 1 . (8a)

For the triangular frequency versus time relationship of figure 1, one can show that the magnitude of the error in eq (8a) is less than

and, for a~10 Hz/Sec, t ' < 100 ns, the effects of this approximation should not

be easily detectable. Equation eq(8) reveals that r(t) is, to a good

approximation, the Fourier transform of h(t) and, from eq (7), we have

The above result may seem intuitively obvious, but nevertheless it is an approximation valid only when w(t) varies slowly enough, and when the effects

of the turning points of the ramp can be ignored.

As a specific example, we will discuss the response of an ideal FM/CW

system to a planar interface located some distance from the antennas. We will assume that the relative delay, t, is small and that the propagation media are not strongly dispersive so that we are well within the range of our previous

approximations.

The physical situation is sketched in figure 4a. The reflection

coefficient, R(u>), is given by

10

r(t) = R [o>( t ) ]

.

(9)

III. 3 Response From a Dielectric Interface

10
R ( oj) =rex p-iwx ( 10 )

where r, the reflection coefficient at the interface, is determined by the

contrast of electrical properties across the interface. The media are said to

be "dielectric" if the loss tangent, a/we, is small (a is the conductivity

and e is the permittivity). Under these conditions, r, may be approximated

as

/i7 - /ej

r

= /ej" + /ejT

(normal incidence). The subscripts 1 and 2 refer to the near and far sides of

the interface, respectively. If e is independent of the frequency, the media

are said to be "nondispersive." Thus, it may be stated that for non- dispersive, dielectric media, r is real and independent of frequency. Note

that r is positive if medium 1 is "more dense" than medium 2, (e^ > e^), and

negative if medium 1 is less dense than medium 2, (e^ < e 2

).

The frequency versus time dependence of figure 2a will now be assumed so that

r(t) = R[w( t)] (9)

will be a periodic function of time. The spectrum of r may be found by

expanding it in a Fourier series:

where

r(t) = rexp-iw x[a /2 + z ( i) n

a cosw t]

,

a o

n =i

n n

2

w

d

sinCU^ - w

n

) P/4]

a

n + w

n L(wj - u

n

) P/4J

( 11 )

( 12 )

In equations (11) and (12): w

a is the average, or midband frequency (see

Ojj XL

figure 2a); w

n = —- n is the n Ln harmonic of the sweep repetition rate; and w

d

is the difference frequency, ax, as defined in eq (1).

The real and imaginary parts of r(t) = A + iB may be written explicitly

as

11
A(t) = rcosu^x [a

Q

/2 + Z (-)

n a 2n cosw

2n

t]

n=l

°° n (13)

- rsinu^x [ n L

( ' ,a

2n-

^COSo)2n _^t]

00 B(t) = - rcosu^x [ z (-)

na

2

^cosuu .t]

n=l

oo - rsinoi t [a /2 + Z (-)

n

a 2nC0Su)2n ^ * n=l

Some basic properties of the spectrum of r(t), which may be deduced from

eqs (11), (12) and (13) are noted below:

(1) The spectrum is discrete, containing contributions only at

frequencies, w , which are harmonics of the sweep repetition rate.

(2) Though the spectrum is peaked at the difference frequency, there are,

2 2 in general, sidebands present which decay as 1 / ( to ).

(3) The coefficients, a

n

, do not depend on the sweep period, P; hence,

there is no theoretical reason to prefer one sweep rate over another

2 (as long as 1/2 ax << 1) .

(4) Only cosine terms appear in the expansion, shown in eq (11). This

will be true for any frequency versus time relationship which is even

in time (i.e., u>(t) = ai(-t)). Such terms will be referred to

loosely as being "in phase". The appearance at any sine

( "quadrature") terms may be attributed to the breakdown of this even symmetry

.

(5) It is possible to adjust the average frequency, w , so that the

a spectrum of A or B (eq (13)) will contain only odd or even harmonics

of the sweep frequency. This will be true, in general, for any time versus frequency relationship which is "antisymmetric" about the

quarter period point (w(t)-u> = -w(P/2-t) + u>_).

d a The more general case of a target which is a multiple layer medium may

also be treated using the analysis of this section. This is the case since

the reflection coefficient for a stratified medium may be broken up into a sum

of terms corresponding to the right hand side of eq (10) [5,6]. The delay for

each term corresponds to the time of transit for one of the various paths from

the transmitting to the receiving antenna. Figure 4b, which is a sketch of a

12
multilayer target, shows several of these paths. The solid line is an example

of a "primary path" — one which involves a single reflection from one of the

interfaces. The dotted line is an example of a "secondary path" -- one which

involves three or more reflections from two or more interfaces. When the

contrast between the various layers is low ( r<< 1 ) , the primary reflections

dominate. In this case the spectrum of r(t) will be, to a good approximation,

a superposition of spectra like that of eq (11), and will be peaked at

frequencies corresponding to the range of each interface.

1 1 1. 4 Closed Loop System

The results of the previous section are expected to hold only qualitatively in an actual implementation. This is so mainly because real antennas

do not radiate plane waves and real targets are not infinite planes. There is one situation, however, in which the severity of these approximations is

greatly reduced. Consider the system shown in figure 1, but with the antennatarget-antenna path replaced by a coaxial line. This will be referred to as a "closed loop" system. The spectrum of r ( t) is given in eq (11) with r=l. The

delay, t, corresponds to difference in length between the reference and signal

lines. Since the fields in a coaxial line are, to a very good approximation,

TEM in nature, theory and experiment should be in quantitative agreement.

Though not of much practical importance in its own right, the closed loop

system is a useful tool in evaluating the performance of an FM/CW radar in a precisely defined condition.

1 1 1. 5 Resolution

As can be seen from eqs (11) and (12), the spectrum of the response

function due to a sharply defined interface does not consist of a single

frequency, corresponding to the delay but, instead, is spread out as a series of harmonics with a sinx/x type envelope which is centered at u^. When

two or more such peaks are superimposed in the spectrum, they will overlap and

may not be distinguishable if the spacing between them is too small. We will

arbitrarily adopt the convention that two peaks are resolvable when they are separated by more than the half-width of the central lobe. (Rayleigh

Criterion) [7].

13
From eq (12), the location of the first zero of the envelope, u> , is

given by

w

r

= 4tt/P ,

which gives a minimum resolvable delay, t , of

= uy/a = tt/Acj = — ns (14)

where we have used the fact that for the triangular ramp of figure 2,

_ 6 a = 4Au)/p. Af = 2Aa)/2n x 10 is the bandwidth in MHz. Thus we would say

that if the differential delay between two paths is less than t , they would

not be resolved as two distinct paths by the FM/CW system. A bandwidth of 1 GHz will produce a resolution of 1 ns, which corresponds to a distance of

about 30 cm//F in a medium of relative dielectric constant, e. The resolution

condition, eq (14), is somewhat arbitrary and serves as a standardized criterion for comparing different systems. In an actual case, an experimenter

might be able to do considerably better, especially if some constraints may be

placed on the nature of what is being probed (e.g., the target might be a uniform slab of some unspecified thickness which is to be determined, etc.).

It is interesting to note that x^ depends only on the bandwidth. Other

system parameters may degrade the signal and reduce the effective resolution,

but the ultimate achievable resolution for any system depends on the bandwidth

alone.

1 1 1. 6 Tuning the FM/CW System

As has been noted previously, the spectrum of the response function,

r(t) , for an isolated dielectric interface will generally contain sidebands.

Under certain conditions, however, it is possible to "tune" the apparatus so that the only frequency present in the spectrum corresponds to the difference

frequency, = ax. An inspection of r(t) (see eq (10)) shows that it is

basically sinusoidal; the sidebands in its spectrum are due to discontinuities

in slope which occur at the turning points in the frequency ramp of figure

1. A typical situation is shown in figure 5. Tuning consists of altering the

system parameters so that these discontinuities disappear.

Either by graphical analysis or by study of the Fourier Series, eq (13), one can show the system is "in tune" when the difference frequency, u^, and

14
the average frequency, w

a

, (see 1 1 1. 3) satisfy the conditions

2tt w

d

' P“

1

w

a = "27 m *

i and m are arbitrary, independent integers. When % and m are both even or

both odd, the response will be in tune at the cosine port. When a is even and

m is odd, or vice versa, the response will be in tune at the sine port.

Notice that the difference frequency will always be a multiple of the sweep

repetition rate, 2n/p, when the system is in tune.

Now consider a constant bandwidth system. (For maximum resolution, it is

good to keep the bandwidth as large as possible.) The tuning conditions may

be written as follows:

tt a 2Aw

(15a)

- m A w = -^-Aw. a l

(15b)

(2a

is the bandwidth).

The system is tuned by first adjusting the delay so that eq (15a) is

satisfied for some l. (The delay is most easily varied by inserting a linestretcher or equivalent device into the signal path.) As viewed on the

spectrum analyzer, this adjustment will cause the spectral amplitude at id = ^

to be maximized, while the amplitudes at the harmonics

+ 2n’

n = W1 ^ be rec(uced t° zero. (A perfect system is

assumed, of course.) Next, the average frequency is adjusted until eq (15b)

is satisfied for some m. On the spectrum analyzer this condition corresponds

to the zeroing of the spectral amplitudes at the harmonics

w £+2n l*

n =

> ±2 > **** when ecl s anCl (15b) are satisfied, of

course, the only remaining harmonic is:

If we choose the sweep range in the right way, it is possible to tune the

system by only varying the delay. For instance, if the sweep range is one octave, w = T 2ao) and condition eq (15b) is automatically satisfied with m =

a C15
3 i. When the bandwidth is one octave, it is possible to tune the cosine port

at every harmonic of the sweep rate by varying the delay alone. The sine port

will never be in tune since with m = 3i, m and l are always both odd or both even. (It is interesting to note that there is no value of the ratio m/i

which allows tuning of the sine port for all harmonics. Tuning will always be

at the cosine port or alternately at the cosine and sine ports.)

Tuning is valuable in two ways: First, in the "closed loop system" the

degree to which you can tune is a direct measure of the quality of the microwave hardware. Second, in the field, tuning can be used to remove the

sidebands of the main surface reflection. These sidebands normally fall off

rather slowly and can obscure smaller signals. It should be emphasized that

it is possible, in general, to tune the response only for one reflector at a

ti me

1 1 1.7 Numerical Examples

This section contains computer simulations of typical FM/CW signals. The

response is assumed to be due to a dielectric interface, r = 1, located a distance, d, away from the system. The spectrum is calculated from eq (13)

r\ J using a delay of t = — (c = speed of light). The amplitudes at the first 30

harmonics are plotted for both the sine and the cosine ports.

Figure 6 is a simulation of an FM/CW system operating over a one octave

range (1-2 GHz). The spectrum of the response function is plotted at several

depths ranging from 0.975 m - 1.06875 m. At a depth of 0.975 m, the system is

in tune; the sole excited harmonic at the cosine port is number 13. As the

depth increases, note the amplitude in the sidebands which might obscure lower

level signals. At depth 1.05 m, the system is again in tune with only the

14th harmonic excited. From the equations of the previous section it may be seen that the system will pass in and out of tune every 0.075 m and that it is

possible to tune only the cosine port. Had a reflection coefficient, r, of -1

been used, all of the patterns would be reversed in polarity. This fact makes

it possible to distinguish between a phase shift of 0° or 180° on reflection

from the interface.

Figure 7 is similar to figure 6 except that the frequency is swept from 1

to 3 GHz. As can be seen, this system is in tune when the depth is 0.45 m and

is periodically in tune every 0.0375 m. This time, however, the "tuned"

condition oscillates between the cosine and sine ports.

16
IV. Instrumentation

The intent in this section is to give a view of the system as it was used

during the testing phases. Parts of the system may be deduced from the

preceding (theoretical) discussion; however, other parts (for example

leveling) were not mentioned, but have been added for reasons discussed

later.

The system shown in figure 8 is described in two parts, microwave and

signal processing.

IV .1 Microwave System

The microwave source is a Yttrium Iron Garnet (YIG) tuned oscillator.

This device exhibits excellent tuning linearity, low noise, and good frequency

stability. Power output at the source is approximately 40 mw. (It varies by

3 dB over the frequency range 1-4 GHz.) The output frequency is controlled by

applying a voltage to the tuning port. The output frequency varies linearly

from 1 to 4 GHz for a zero to 10 volt tuning range.

As mentioned previously (theory section) the triangular wave modulating

signal is used because of its linearity with time (neglecting the turning

points). While having an oscillator which follows its input voltage is

important, the input voltage linearity is also important. The slope of the

triangular wave must be uniform and must be the same magnitude for both the

positive and negative going portions of the triangular wave. The circuit

shown in figure 9 was designed to achieve those goals. In this circuit the

triangular wave (and synchronous square wave) is generated from a repetitive

pulse source by counting the pulses and driving a digital to analog converter

(DAC) from the binary latched output of a 3 stage up-down counter. The

positive and negative slope portions of the triangular wave are each generated

from 4096 individual pulses. The DAC output is then low pass filtered to remove the 4096 small staircase steps in the up and down ramps of the

triangular wave. A square wave signal synchronous with the triangular wave is

available also and is used for gating in the signal processing system

(described later in this report).

17
Other parts of the microwave circuit are shown in figure 8. The transmit

and receive antennas are ridged waveguide antennas usable in the frequency

range 750 MHz to 12 GHz. At 1 GHz the beamwidth is approximately 35 degrees.

The dividers are 3 dB in-phase power dividers with a frequency range of 1 to 4 GHz.

The mixer is a double balanced mixer usable from 1 to 4 GHZ. This mixer

was selected for the best response (low noise and uniform mixing) from about

ten other mixers from different manufacturers and construction.

The level control part of the microwave circuit includes a pin diode

attenuator, amplifier, and detector. The pin diode attenuator is a voltage

controlled attenuator operating from 1 to 2 GHz. When using the level control

circuit the system frequency is reduced to 1 to 2 GHz instead of from 1 to

4 GHZ. However, the noise at the mixer output is reduced when the leveling

control is in the circuit. The tradeoff is getting a better signal to noise

ratio using power leveling with a reduced bandwidth or getting better

resolution with the full 1 to 4 GHz bandwidth. Both types of operation are available, but not at the same time because a wideband pin diode attenuator

could not be obtained.

The detector samples the power level at the output of the second

divider. That power level is converted to a dc voltage and compared against a reference voltage. The voltage difference between the reference voltage and

detected voltage is amplified and used to control the pin diode attenuator,

thus maintaining a constant output power level.

IV. 2 Signal Processing System

The output from the mixer is a complex signal with major frequency

components of less than 10 kHz. The spectrum analyzer is an instrument

designed to display a complex signal in its basic frequency components. This

is accomplished by digitally sampling the complex input signal and then

converting the digital data to the frequency spectrum using a computer

algorithm called the Fast Fourier Transform (FFT). Sampling is a process of

recording the amplitude of the input signal during a narrow aperture of time

(1 us) and converting that value to a 12 bit binary number. The samples are taken at the rate of 104,000 per second with uniform time spacing between

samples. 1024 samples are recorded for each FFT calculation so the sampling

18
process requires about 10 milliseconds. The FFT calculation can be completed

in less than 0.5 second. Hence, frequency spectrum data can be updated about

every second.

Getting those data into a form useful for indicating the thickness of a layer of coal requires more manipulation. The frequency spectrum data is

transferred to a computer that calculates the thickness by searching through

the frequency data array to sort out the maximum responses. The frequency and

magnitude of the largest response are interpreted to be the position and

magnitude of the air-coal interface. The frequency of the next largest

response following the air-coal frequency is taken as the position of the

coal-shale interface. The thickness of the coal is then calculated from the

frequency difference between those two responses. A correction factor is also

calculated in the computer and applied to the thickness reading. The

correction is due to the change in velocity of propagation for the microwave

signal through the coal. If the dielectric constant of the coal changes, the

velocity of propagation changes also in inverse proportion to the square root

of the dielectric constant. Therefore, the thickness reading which is a function of the transit time for the microwave signal through the coal is

corrected. Two types of correction have been used. In one, the dielectric

constant is assumed and the thickness reading is calculated using that

value. In the second (tracking mode) the dielectric constant is calculated

using a relationship based on the magnitude of the reflection from the aircoal interface.

Following the thickness calculation and correction, the results are displayed on the computer CRT. They may also be printed as shown in figure 10

for a permanent record or stored on magnetic tape.

V. Test Facility

The test facility was designed and constructed to provide a convenient

test area for the CID system. The main advantages of such a test area are accessibility and well defined targets.

For this project, testing is a major task in both the developmental stage

and in demonstration of the final model. If all the testing were performed

underground in coal mines the task would be almost impossible since suitable

testing areas are not easily found and the associated problems of getting the

19
equipment in and out of the mines without disturbing normal mining operations

are very difficult. Also, many mine operators seem reluctant to allow testing

in their mines because of the disruption and the need for some of their own workers to accompany the testing crews, and the safety problems inherent when

inexperienced personnel are underground.

Verification of the target area is another problem in an underground mine

because the coal thickness at the test areas must be measured independently

and correlated with the CID measurements. Additionally, very little

information about the dielectric constant of the coal is available or obtainable independently from the CID measurements.

For these reasons, the test facility was constructed at NBS. However, it was designed so that it could be dismantled and reassembled at another site to

be designated by DOE following the testing at NBS.

A photograph of the test facility is shown in figure 11. The main

structural supports and cross members are made from fiberglass material to

reduce extraneous reflections which would interfere with the CID tests. The

only metallic parts are motors and gears located high in the overhead

structure well away from the testing area. The facility provides movement of the CID system (microwave part) over a 244 cm (8 ft.) by 122 cm (4 ft.) horizontal area (x, y coordinates). Vertical

movement (z) is approximately 76 cm (30 inches). An additional movement, ( e )

,

is provided to tilt the CID antennas plus or minus 45 degrees from normal

incidence to the hori zonal plane.

Positioning the CID antennas over the test area is accomplished by

driving stepping motors for each of the four directions x, y, z, and 9. The

stepping motors are driven from pulse sources that are sequentially applied to

the windings in the motors to provide either clockwise or counterclockwise

rotation of the motor shafts. Gearing was designed to provide movement of

0.00254 cm (0.001 in.) in both x and y directions. Incremental positioning in

z (vertical) is one-fourth of that because of the higher gear ratio needed to

lift the total package. The tilt angle was designed to provide 0.001 degree

of movement for each pulse to the stepping motor. Resetability in each

direction is reduced, however, by a factor of 10 because of backlash in the

gear and motor mechanisms. Estimated requirements of resetability for testing

the CID system were plus or minus V4 cm (0.1 in.) which is well within the

design of the system including backlash.

20
VI. Target Material

The target material used in the test facility was designed to simulate

coal in the frequency range 1 to 4 GHz. The design criteria were relative

dielectric constant and attenuation in that frequency range. Earlier

measurements (Ellerbruch, Belsher, and Adams) had found relative dielectric

constants of coal in the range from 4 to 10. Values for the overlying shale

were always higher (usually 10 to 30). In general, there was a distinct

difference at the coal -shale interface which produced a reflection of the

microwave energy. Attenuation values were approximately 40 dB/m (1 dB per

inch) at 1 GHz. For each octave increase in frequency the attenuation

doubled.

A view of some of the target material can be seen in figure 11 (test

facility). Sixteen of the blocks of material were produced with a nominal

dielectric constant of 4 and another sixteen blocks were produced with a dielectric constant of 9. For each group of sixteen, four have a thickness of

5 cm (2 in.), four have a thickness of 10 cm, and eight have a thickness of

15 cm (6 in.). All the blocks are 61 cm (2 ft.) long by 61 cm wide. The

surfaces were made as flat as possible and either parallel or orthogonal to one another for stacking purposes. The objective in stacking is to get as many combinations of thicknesses between 5 cm and 40 cm as possible without

getting air gaps between the blocks.

The dielectric constant and loss tangent of each block of material were measured by the manufacturer at 3 GHz and by NBS over the frequency range

1 to 4 GHz. The NBS results are given in Table I. These values agree with

the manufacturer's results within the uncertainty of the manufacturer's and

NBS measurements.

VII. Testing

The purposes of the above ground testing program were to optimize the CID

during development and to verify system performance relative to the following

criteria:

(1) acquiring and tracking the coal -shale interface through coal layers

up to 38.1 cm (15 in.) in thickness;

21
(2) correcting the measured thickness for changes in the coal layer

dielectric constant;

(3) tracking the interface independent of the antenna to air-coal

interface distance for variations up to 91 cm (3 ft.); and

(4) acquiring and tracking the coal -shale interface in the environment of

the metal mass of a mining machine.

VI 1. 1 Tracking a Changing Dielectric

Over 300 individual measurements were made on the target material of

varying thickness and different dielectrics. For each measurement the

thickness was displayed using the automatic dielectric tracking mode and also

a nontracking mode. Both results for each measurement were displayed and

printed. Those results were then compared with the known thickness of the

target material. A list of the results from those measurements is given in

Table II. The number of observations refers to the total number of

measurments made at a particular thickness. All measurements of a given

thickness and dielectric constant were not made on the same block or set of

blocks or at a particular antenna height. Therefore, some contribution to

measurement error may be attributed to antenna height change. In general, the

standard deviation of the measurement results calculated from the nontracking

measurements are lower than the measurement results from the dielectric

tracking mode. Also, the agreement of the mean of the nontracking results is

closer to the actual thickness of the material that was measured.

VII. 2 Measuring Through Coal Thicknesses Up To 38 cm (15 in)

The results listed in Table II include measurements at thicknesses

greater that 38 cm (15 in.). These results are not significantly worse than

other measurements made at smaller thicknesses. The target material was made

to simulate coal in attenuation as well as dielectric constant. Four types of

measurements were attempted: (1) K4 material over a metal plate, (2) K4

material over K9 material, (3) K9 material over a metal plate, and (4) K9

material over K4 material.

The back surface could be identified in all but type (4) tests through

40 cm (16 in.) of simulated coal. Measurement (2) is presumed to be the most

representati ve of the measurement in an actual coal mine.

22
Table 1. Dielectric Constant and Loss Tangent of Simulated Coal Target

Material in the Frequency Range 1 to 4 GHz

Identification Thickness (cm) Dielectric Constant Loss Tangent

NBS 15 5.08 4.52 0.08

NBS 8 5.08 4.12 0.07

NBS 29 5.08 4.29 0.06

NBS 28 5.08 3.82 0.07

NBS 12B 5.08 8.44 0.40

PER 13 5.08 7.61 0.14

PER 14 5.08 6.90 0.12

NBS 3 5.08 6.88 0.12

PJC-14P-2 10.16 7.45 0.12

PJC-14P-4-12 10.16 8.73 0.16

PJC-14P-5 10.16 7.78 0.09

PJC-14P-8 10.16 7.44 0.09

NBS- 12 10.16 3.94 0.07

NBS-14 10.16 3.70 0.06

NBS-17 10.16 3.71 0.06

NBS-6 10.16 4.10 0.07

NBS-7 15.24 3.84 0.05

NBS-21 15.24 4.47 0.05

NBS- 16 15.24 3.98 0.04

NBS- 13 15.24 3.91 0.06

NBS- 1

1

15.24 3.65 0.05

NBS-9 15.24 3.68 0.04

NBS-18 15.24 3.45 0.03

NBS-23 15.24 4.65 0.06

NBS-Red-4 15.24 8.73 0.12

NBS-Red-6 15.24 8.91 0.13

NBS-Red-1 15.24 8.72 0.13

NBS-Red-7 15.24 9.25 0.13

PJC-14P-14 15.24 7.17 0.07

PJC-14P-11 15.24 8.80 0.15

NBS-Red-5 15.24 8.28 0.14

NBS-Red-3 15.24 8.66 0.12

23
Table II. Thickness Measurement Results Using The CID

System and Simulated Coal Target Material

Measured

Thickness

Thickness Tracking Standard

Materi al (cm) D i el ectric Devi ation

K4 5.08 3.96 2.67

K4 10.16 9.30 5.05

K4 15.24 10.01 4.88

K4 20.32 14.83 5.94

K4 25.40 24.82 4.72

K4 30.48 21.11 4.24

K4 35.56 30.28 7.44

K4 40.64 36.65 8.10

K9 5.08 4.55 3.58

K9 10.16 9.25 4.90

K9 15.24 9.53 4.75

K9 20.32 16.64 5.69

K9 25.40 25.15 8.74

K9 30.48 19.13 7.01

K9 35.56 27.33 6.32

Measured

Thickness

Not

Tracking

Standard

Deviation

Number

of

Observations

14.58 10.01 37

13.26 4.90 37

22.63 13.82 24

19.66 4.98 11

26.49 1.80 10

29.44 0.23 18

36.91 4.37 26

41.35 5.33 18

7.04 3.30 16

12.45 4.85 20

17.12 6.27 17

19.99 3.94 17

26.80 4.98 20

29.03 6.12 16

34.49 3.00 20

24
VII. 3 Tracking the Interface Independent of Height

As mentioned in Section VII. 1 all the measurements used to tabulate the

results in Table II were made at different heights within the range 65 cm

(26 in.) to 110 cm (44 in.). Part of the error can be attributed to antenna

height change.

Another test was performed to show antenna to coal distance effects.

Those results are shown in figure 12. In this test the thickness was obtained

using only the dielectric tracking mode. A system calibration from a flat

metal plate was obtained at the first distance (shortest). All subsequent

measurements made at longer distances were calculated with reference to the

calibration data. This was done because that method of use is most likely to

be the method used in an actual working site. That is, if the system were

used in a mine, the calibration (used as a reference for the dielectric

tracking mode) would be performed only one or two times per hour or less.

Therefore, measurements made at arbitrary distances from the air-coal surface

would be referenced to a calibration that was probably performed at a different distance. If there was no effect from varying distance then a different calibration distance would not be noticed.

These results show a definite increase of error when the measurement

distance becomes large. Errors from other normal measurements made where the

calibration distance was near that of the measurement were smaller. The

results shown in Table II were, in general, made somewhere near the

calibration distance.

The results shown in figure 12 also show an oscillation with distance

which is to be expected from the discussion given in section 1 1 1. 6 (Tuning the

FM/CW System). That is, the system goes in and out of tune as the distance

from the antenna to the reflecting interface changes. The oscillations can be

reduced by tuning the system for a maximum response at each measurement

point. However, if the CID system requires tuning for each measurement it can not be of much practical use in an actual coal mine measurement situation. A

goal of this project was to develop a system that would give correct answers

with a minimum amount of operator involvement so that the system could

eventually be used on a continuous basis - presumably with an automated mining

system.

25
Tuning the FM/CW system is accomplished by setting the start, stop, and

average frequency of the microwave source. This requires adjustment of the

modulating triangular wave. These are manual adjustments and can be done in

several minutes, but the procedure is too cumbersome to be worthwhile for a system that is intended to be used on a continuous basis. Therefore, the test

procedures did not include tuning.

VI I. 4 Acquiring and Tracking the Coal -Shale Interface

in the Environment of Metal Surfaces

This test was performed to simulate conditions that a coal thickness

sensor would encounter in actual mining situations near large masses of metal

that might produce extraneous reflections.

For this test, metallic wings were attached to the CID system. These

wings extended horizontally away from the antennas approximately 60 cm in two

directions. With these wings in place normal tests were run to measure four

different thicknesses of simulated coal. No discernible effect from the metal

surface was noted in either the operation of the CID system or the results.

Those results are shown in figure 13. The material was arranged as shown by

the solid line. Measurements were made over the mid-points of each block of

material and the thickness was recorded for both the tracking and non-tracking

modes of operation. The antenna height was kept constant for each

measurement.

VIII. MSHA Approval and In-Mine Tests

Obtaining an MSHA experimental permit to use the CID system near the

cutting areas of a coal mine was a task for this project. That permit was necessary for the final phase of in-mine testing where the CID system was to

be tested on an underground mining machine. Figure 14 illustrates in block

form the system submitted to MSHA for approval. All the circuits shown were

to be located inside an already approved explosion proof enclosure. The

spectrum analyzer and computer were to be located out by the first crosscut on the fresh air side of the cutting area.

A decision to discontinue underground testing efforts was made following

the above ground tests. That decision removed the need for an experimental

26
permit and the experimental permit request was cancelled in October 1981 with

a letter to MSHA from DOE.

IX. Project Conclusion

The above ground testing of the CID system was completed with a demonstration test in August 1981. A decision based on those tests was then

made not to continue with further efforts towards underground testing and to

devote the remaining time toward project documentation and delivery of the

equipment to DOE. That decision was based on several factors. One was that

the system was too sensitive to antenna to air-coal distances which made

continuous operation difficult. Another factor was that another type roofcoal monitoring system using natural background radiation was already in use with accuracies acceptable to DOE. Further efforts towards improving and

testing the FM/CW type CID were not considered worthwhile by DOE.

Acknowl edgments

Thanks are extended to Robert Metzker and Douglas Tamura for their

efforts in this project. Mr. Metzker supervised work and provided liaison

with the NBS instrument shop to develop the above ground test facility.

Mr. Tamura did most of the packaging and testing of the CID system.

Mr. Victor Lecinski of the NBS instrument shop designed the test facility

and Mr. Joe Schneider did the machine work.

Mr. Doyle Ellerbruch provided background information and advice

throughout the project.

27
X. References

[1] Ellerbruch, Doyle A. and John W. Adams, Microwave Measurement of Coal

Layer Thickness, U.S. Department of Commerce, National Bureau of

Standards NBSIR 74-387 (September 1974).

[2] Ellerbruch, Doyle A. and Donald R. Belsher, FM-CW Electromagnetic

Technique of Measuring Coal Layer Thickness, U.S. Department of Commerce,

National Bureau of Standards NBSIR 76-840 (May 1976).

[3] Oppenheim, A.V. and Shafer, R.W., Digital Signal Processing, (PrenticeHall, Inc.), pp. 358-361, Englewood Cliffs, NJ (1975).

[4] Estin, A.J . and Daywitt, W.C., Evaluation of Signal Plus Noise Detection

Error in an Envelope Detector with Logarithmic Compression, U.S.

Department of Commerce, National Bureau of Standards NBSIR 79-1614

(1979).

[5] Lytle, R.J. and D.L. Lager, Using the Natural -Frequency Concept in Remote

Probing of the Earth, Radio Science, JJ_, No. 3, pp. 199-209 (Mar. 1976).

[6] Mathews, J. and R. L. Walker, Mathematical Methods of Physics, (W.A.

Benjamin, Inc.), pg. 102, New York, NY (1970).

[7] Tretter S.A., Discrete-Time Signal Processing, (John Wiley and Sons, New

York), pp. 21-22, 230-231, 291-222 New York, NY (1976).

[8] Wittmann, R.C., Simulated Pulse Techniques for Measuring Coal Layer

Thickness, Chapter 6, pp. 178-297. Analytic bases for electromagnetic

sensing of coal properties, Final technical report (July 1978).

U.S.D.O.E. , ET-75-G-01-8972.

28
TARGET

Figure 1* A simple FM/CW radar block diagram. "D‘ denotes a divider, M

denotes a mixer.

29
>-u

z

Ui

D

0

Ui

oc

Figure 2a. A triangular frequency versus time relationship useful in FM/CW

radar applications.

Figure 2b. The reference signal and the target signal are displaced in time by an interval, x, which is the difference in travel times for the two

paths of figure 1. The displacement in time corresponds to a displacement of frequency, u).. The FM/CW system detects this

"beat" frequency.

30
Figure 3

SIN<£>

SINE PORT COSINE PORT

. Use of a phase discriminator for the detection of the real and

imaginary parts of r(t).

31
€,,G.

Figure 4a. Reflection from an infinite plane boundary between two media.

Figure 4b. A primary ray path (solid line) and a secondary ray path (dotted

line). Secondary paths result from multiple reflections within the

medium.

32
FREQUENCY

Figure 5. This figure shows a hypothetical response, r(t), from a planar

interface. r(t) is basically sinusoidal, but the slope is

discontinuous where the frequency ramp changes slope. These

discontinuities contribute sidebands to the spectrum of r(t).

33
Figure 6.

COSINE PORT SINE PORT

Depth = 0.97500 m.

j i

i

\

;

i

i

i

i

l

i

;

i

\ re re | • re

I

i

re re

Depth = 0.99375 m.

j

, 1 .

1 ,

1 . 1 , 1

j

•

| *

i

•

» *

Depth = 1.01250 m.

1 1 1 1 1 1 1 1 1

1

Ml

i

1 m

1 • • • 1 1

1

1

in

re

•

i •

i

*

i

*

i

1

1

1

j

I Ij'i'i

re

Simulation of an FM/CW system responding to an interface at various

distances. The sweep range is from 1-2 GHz. Shown is the spectral amplitude

at each of the first 30 harmonics.

34
COSINE PORT SINE PORT

Depth * l.(

»

1

l

D3125 m.

i

r

j

| .l.l.l.l.l.i,

i

i

i

,1

< M 1 1 1 1 1 1 1 1 I 1 1 i"

j

i

!

i n m »

( Oepth = l.(

1

j

|

I

l

!ii J o8 E

—

-

— .l.l.l.l.l.i.

1

i

i

j

Il

j

i

I n a ~ =

• -j*r i-i*

n

Depth = 1.06875 m.

1

{

j vrrr i'i'j

• 1 1 1 1 1 1 1 1 1 1 i ! i |M

- 9 » a n

n

Figure 6. (continued)

35
SINE PORT

Figure 7.

COSINE PORT

Depth = 0

i

i

»

i

i

/

i

.45000 m.

i.i ill

!

i

)

i

j

i

I «

1

• i

1

j — ss » m ! “ —

*

1

*

i

*

i

*

•

*

|

1

i a m j

i

iDepth = 0

J

s

J

>

1

.46875 m.

r.I.I.I.I.I.

i

j

i

i

|

i

i

i

1 *rriT|T

i *

i

i

:

i

j “ a

l - . -

t

j

j

i

i n m j a

;

T

n

;

I

1

]

I a m

Depth = 0

i

i

i

i

i

.4B750 m.

i4

i

i<

j -"Hi

i

i • a a m 11 C K !. ii

.

...

i

Simulation of an FM/CW system responding to an interface at various

distances. The sweep range is from 1-3 GHz. Shown is the spectral amplitude

at each of the first 30 harmonics.

36
COSINE PORT SINE PORT

Depth = 0.50625 m. 1

I tit J|*

I . I . I . I . I . • . •

.

• *

I

- *

*

I

* I

• I

.1.1.1.

Depth = 0.52500 m.

riTri’l

.l.l.l.i.i. ......

Depth = 0.54375 m.

,i. i.i.i.I.I.

J

* l

*

* • *

i

* I

*

I *

!

* |

' |

. l . . . i . *

.

Figure 7. (continued)

37
SAMLC

COAL

A/R

~7T

-h+

- \ T~

// ' v //

'/

\ \

' \W

Figure 8. Block diagram of the FM/CW system used as a coal interface

detector.

38
Figure 9, Schematic circuit of the triangular wave generator

used in the FM/CW system,

39

COj
The plate response Is: The surface response Is: This corresponds to a target relative

dielectric constant of:

The difference frequency corresponding

back surface of the target Is: The delay Is:

-23.44 dB at 66 cm -30.69 dB at 80 cm

6.41

to the delay from front surface to 75.00 Hz 3.41 nS

The thickness, calculated with the measured value of the relative dielectric constant Is: 20.2 cm The thickness (not corrected) Is: 20.9 cm

Dielectric constant: 6.00 Antenna height: 66 cm Thickness In centimeters: 20.3 Block number: NBS 23 4 8

CH R: - 20dBV FS lBdB^DIV

MKR: - 47.4dBV

Figure 10. A copy of the computer results and display from a CID measurement.

40
Figure 11. Photograph of the CID test bed facility.

41
(NO) SS 3NXOIH1 Q3dnSV3W

42

Figure

12.

Effect

of

antenna

to

interface

distance

on

thickness

measurements
K

9

MATERIAL

t

U

•

0

ui

H

u

ui

tt

ec0

u

0

ui

u

UJ

tc

tt0

u

H

0

z

1

•

z

0

550

A

t

r

in

"I I i i i I I |

i r o m o « N

(WO) 1HDI3H

~m—

r

43

Figure

13.

Results

of

measurements

made

in

the

presence

of

metallic

surfaces.

Solid

line

represents

the

top

surface

of

the

dielectric

material

.
L J

Figure 14. Block diagram of the FM/CW system proposed for an experimental

permit. The power cable and fiber optic cable (50 meters long) go

to the spectrum analyzer and power source located in a non- hazardous area of the mine.

44
U.S. DEPT. OF COMM. 1. PUBLICATION OR 2. Performing Organ. Report No. 3. Publ ication Date BIBLIOGRAPHIC DATA

SHEET (See instructions)

REPORT NO. NBSIR 82-1663 May 1982

4. TITLE AND SUBTITLE

Improved Coal Interface Detector

5. AUTHOR(S)

Keith C. Roe and Ronald C. Wittmann

6. PERFORMING ORGANIZATION (If joint or other than N BS. see instructions)

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

WASHINGTON, D.C. 20234

7 pggn tract/Grant No. ET-77-C-01 -8881

8 . Type of Report & Period Covered

9. SPONSORING ORGANIZATION NAME AND COMPLETE ADDRESS (Street. City, State. ZIP)

Department of Energy

Pittsburgh Mining Technology Center

Pittsburgh, PA

10.

SUPPLEMENTARY NOTES

| | Document describes a computer program; SF-185, FIPS Software Summary, is attached.

11.

ABSTRACT (A 200-word or less factual summary of most significant information. If document includes a significant

bibliography or literature survey, mention it here)

This report describes the theory, design, construction and testing of an electromagnetic coal interface detector. The purpose of this type detector is measuring the

thickness of roof coal left during underground mining operations. An above ground

test facility constructed to evaluate the coal interface detector is also described.

12.

KEY WORDS (Six to twelve entries; alphabetical order; capitalize only proper names; and separate key words by semicolons)

Coal-shale interface; electromagnetic coal interface detector; FM/CW radar;

roof-coal thickness

13.

AVAILABILITY

Unlimited

1 | For Official Distribution. Do Not Release to NTIS

| Order From Superintendent of Documents, U.S. Government Printing Office, Washington, D.C. 20402.

[X] Order From National Technical Information Service (NTIS), Springfield, VA. 22161

14. NO. OF

PRINTED PAGES

52

15. Price

$9.00

USCOMM-DC 6043-P80

U.S. GOVERNMENT PRINTING OFFICE: 1982-579-119/295 REGION No. 8