NTIA Report 97-338

Interference Simulation for Personal

Communications Services Testing,

Evaluation, and Modeling

James G. Ferranto

U.S. DEPARTMENT OF COMMERCE

William M. Daley, Secretary

Larry Irving, Assistant Secretary

for Communications and Information

July 1997

PRODUCT DISCLAIMER

Certain commercial equipment, instruments, or materials are identified in this paper to specify

adequately the technical aspects of the reported results. In no case does such identification imply

recommendation or endorsement by the National Telecommunications and Information

Administration, nor does it imply that the material or equipment identified is necessarily the best

available for the purpose.

iii

CONTENTS

Abstract ..............................................1

1. INTRODUCTION ......................................1

1.1 PCS Testing, Modeling, and Evaluation Program Simulation Methodology.......2

1.1.1 Radio Link Simulation (Physical Layer) ......................3

1.1.2 Bit Error Simulation (Physical Layer) .......................4

1.1.3 Network-levelParameterComputationandNetworkSimulation(DataLinkLayerandAbove) 4

1.2 Noise/Interference Modeling and Channel Simulation..................5

2. NEEDFORNOISEANDINTERFERENCEMODELINGINTHE2-GHZPCSENVIRONMENT . . 7

2.1 Interference Environment ................................7

2.1.1 Statistical Interference Models...........................8

2.1.2 Standards Requirements for Interference Levels in PCS Systems .........9

2.1.3 System-specific Models and Interference Simulation . . . . . . . . . . . . . . 10

2.2 Hardware Simulation of PCS Channels in the Noise/Interference Environment . . . . 10

2.3 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

3. PCS CELLULAR GEOMETRY FOR INTRASYSTEM COCHANNEL INTERFERENCE 13

3.1 Assumptions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

3.2 Interference Waveform Notation . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

3.3 Intercell Uplink Interference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

3.3.1 Uplink Cell Geometry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

3.3.2 Adjacent Cell Distances . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

3.3.3 Second-level Cell Distances . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

3.4 Intercell Downlink Interference . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

3.4.1 Downlink Cell Geometry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

3.4.2 Adjacent Cell Distances . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

3.4.3 Second-level Cell Distances . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

3.5 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

4. GLOBAL SYSTEM FOR MOBILE-BASED PCS 1900 INTERFERENCE WAVEFORM 25

4.1 PCS 1900 Physical-layer Overview. . . . . . . . . . . . . . . . . . . . . . . . . . . 25

4.2 Theoretical Definition for PCS 1900-based Gaussian Minimum-Shift Keying . . . . . 26

4.3 PCS 1900 Modulated Waveform Generation . . . . . . . . . . . . . . . . . . . . . . 28

4.3.1 Time-domain Representation of the Phase Expression . . . . . . . . . . . . . . 29

4.3.2 Calculation of the PCS 1900-GMSK Phase Pulse Using Numerical Integration . 32

4.4 Power Considerations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

4.5 Timing and Synchronization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36

v
4.6 Interference Expressions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36

4.6.1 Uplink Interference Expressions . . . . . . . . . . . . . . . . . . . . . . . . . 37

4.6.2 Downlink Interference Expressions . . . . . . . . . . . . . . . . . . . . . . . 39

4.7 Computer Simulation of the PCS 1900 Noise and Interference Environment. . . . . . 40

4.7.1 Uplink Simulation Methodology for Noise and Interference Generation . . . . . 41

4.7.2 Example Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44

5. IS-95-BASED CDMA-PCS INTERFERENCE WAVEFORM . . . . . . . . . . . . . . . 53

5.1 CDMA-PCS Physical-layer Overview . . . . . . . . . . . . . . . . . . . . . . . . . 53

5.1.1 Reverse Link Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53

5.1.2 Forward Link Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 55

5.2 Offset QPSK Waveform Expression (Reverse Link) . . . . . . . . . . . . . . . . . . 57

5.2.1 Markov Chain Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59

5.2.2 I and Q Explicit Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62

5.3 QPSK Waveform Expression (Forward Link) . . . . . . . . . . . . . . . . . . . . . 63

5.4 CDMA-PCS Power Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65

5.5 Timing and Synchronization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66

5.6 Interference Expressions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 67

5.6.1 Uplink Interference Expressions . . . . . . . . . . . . . . . . . . . . . . . . . 68

5.6.2 Downlink Interference Expressions . . . . . . . . . . . . . . . . . . . . . . . 70

5.7 Computer Simulation of the CDMA-PCS Noise/Interference Environment . . . . . . 73

5.7.1 Uplink Noise/Interference Simulation . . . . . . . . . . . . . . . . . . . . . . 74

5.7.2 Example Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 77

6. SUMMARY AND CONCLUSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . 85

7. REFERENCES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87

APPENDIX A: MODULATED WAVEFORM NOTATION . . . . . . . . . . . . . . . . . 89

APPENDIX B: COMPLEX BASEBAND NOTATION . . . . . . . . . . . . . . . . . . . . 91

APPENDIX C: ACRONYMS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 95

vi
INTERFERENCE SIMULATION FOR PERSONAL COMMUNICATIONS SERVICES

TESTING, EVALUATION, AND MODELING

James G. Ferranto1

Abstract

An interference model applicable to wireless technologies is presented in this

report. Specifically, a generic methodology for cellular system self-interference

modeling was developed, then applied to two proposed personal communications

services (PCS) technologies: the Global System for Mobile-based PCS 1900, and

IS-95-based code division multiple access. Resulting system-specific models are

discussed in detail, and are used to produce output noise and interference

waveforms suitable for implementation in a real-time hardware channel simulator,

or as a component of a higher-level software simulation. Example outputs are given

for simulations of both technologies, with corresponding statistical analyses of the

noise and interference waveform properties. Models described in this report are

particularly well-suited for independent PCS system evaluation by other Federal

agencies, system manufacturers, and service providers.

Key words: noise; interference; personal communications services; simulation;

model

1. INTRODUCTION

Widespread implementation of personal communication services (PCS) is expected to

revolutionize telecommunications in the United States within the next few years. The digital nature

of PCS, in conjunction with high channel reusability, allows a variety of user services in dense

coverage areas. Commercial and residential PCS services include near-wireline quality voice and

low-speed data in a mobile environment. Extensive control channel designs allow extended

mobility management, roaming, security, and priority access services not previously available in

analog cellular systems.

A combination of low-cost, wide availability, and extended services makes PCS a highly desirable

collection of services for Federal wireless requirements.2 Consequently, the Federal government

plans on making extensive use of commercial PCS where practical. However, certain Federal users

1 The author was with the Institute for Telecommunication Sciences, National

Telecommunications and Information Administration, U.S. Department of Commerce,

Boulder, CO 80303 when this work was done.

2 The Federal Wireless Policy Committee has summarized Federal wireless requirements in a

document entitled “Current and Future Requirements for Federal Wireless Services in the

United States,” which has been presented to relevant wireless standards organizations. In

addition, the United States Government is currently developing a Government-wide

procurement program for wireless services and devices.
mandate specific additional services that must operate acceptably in the PCS environment. For

example, special Federal requirements include priority access and channel assignment (PACA) in

both nonemergency and emergency wireless applications, and enhanced security services provided

by STU-III telephony. These special applications may impose additional performance and

interoperability requirements that are specific to the particular PCS technology, and to the

corresponding deployment.

Even without considering special Federal user requirements, a strong need for PCS evaluation

methods is apparent. PCS systems are largely untested in actual deployment scenarios. Because

predeployment field testing of all aspects of an actual PCS system is prohibitively time consuming

and expensive, alternative methods for evaluation are required. To aid assessment of proposed PCS

technologies, the Institute for Telecommunication Sciences (ITS) has developed synergistic

programs for PCS network testing, modeling, and evaluation. Beneficiaries of these programs include

other Federal agencies, wireless service providers (especially resource-limited service providers), and

wireless equipment manufacturers. Outputs include quantitative and qualitative performance metrics,

interoperability studies, and scenario plans for present and future PCS system operation.

Essential elements for accurate PCS network testing, modeling, and evaluation are system-specific

interference models and their corresponding hardware implementations. Interference model

development is detailed in this report. First, a brief overview of the ITS simulation methodology is

provided for perspective.

1.1 PCS Testing, Modeling, and Evaluation Program Simulation Methodology

ITS’ expertise in both physical-layer and network-layer modeling provides a unique in-house

interlayer analysis methodology. Work structure for PCS system modeling is divided by layer in a

hierarchical manner: the results of each layer analysis provide the parameters for adjacent layers.

Abstraction is used to simplify higher-layer models, and to reduce computational complexity which

often makes a model impractical. First, the physical-layer analysis data is supplied to the link-level

analysis; this drives the network-layer analysis, and so on. Conversely, certain parameters in the

lower layers are set based on higher-layer conditions. For example, channel self-interference

(a physical-layer issue) in a code-division multiple access (CDMA) system is proportional to the

number of current active links (a network-layer issue, Figure 1.1).

2
1.1.1 Radio Link Simulation (Physical Layer)

The radio link analysis encompasses all physical effects on a PCS mobile station/base station pair.

Three major components, shown as the top three boxes in Figure 1.1, provide the description of the

radio link environment:

1. Propagation Model: The propagation model is developed from both theoretical

derivation and observed data. The propagation model describes effects of the

physical channel on the communication waveform. Attenuation, multipath, and

Doppler shift profiles are all components of the propagation model.

2. Noise and Interference Model: This model includes all extra-waveform

impediments to the proposed PCS system. The noise and interference waveform

3

Noise and

Interference

Model

Networklevel

Simulation

Bit Error

Simulation

Radio Link

Simulation

Radio

Model

Propagation

Model

Network-level Parameter

Computation

Figure 1.1. Layered approach to wireless simulation.
model is comprised of three components: 1) complex, modulated, zero-mean

Gaussian noise; 2) interference from other users of the PCS frequency band; and 3)

impulsive artificial and natural noise.

3. Radio Simulation: Radio simulation includes all effects of the radio

receiver/transmitter design on communication. Modulators, demodulators,

encoders, and antennas are all part of the receiver/transmitter design.

1.1.2 Bit Error Simulation (Physical Layer)

A statistical model that allows mapping of the major link variables into parameters of a statistical

distribution is derived from detailed bit error information obtained from the radio link simulation.

This statistical distribution is a function of all the link variables, including the channel, modem,

coding, equalization, signal-to-noise ratio (SNR), and all interference. Interference is generated

using the techniques described within this report. The output of the bit error simulation is a bit error

mask that provides statistics for the network-layer analysis.

1.1.3 Network-level Parameter Computation and Network Simulation (Data Link Layer and Above)

The bit error mask created by the bit error simulation is used as input data for a systemwide PCS

network simulation. Using discrete event simulation tools and abstraction techniques, ITS is

capable of simulating a complete PCS system. First, the statistical bit error model is used to profile

packet or frame error statistics (this decreases simulation time significantly, which is an optional

but highly desirable step). The discrete event simulator is then used to model the PCS system, or a

particular component of the PCS system. Any level of system detail is possible; the only limitations

are simulation and development time for the model. The following issues may be addressed in the

network simulation:

• interoperability,

• security,

• protocol analysis,

• mobility,

• priority access,

• channel sharing, and

• deployment planning.

4
1.2 Noise/Interference Modeling and Channel Simulation

An important consideration in the assessment of proposed PCS systems is acceptable performance

in an RF noise and interference (N/I) environment. Service quality on all layers of communication

are affected by the nature of the N/I, with some applications more susceptible to certain types of N/I

than others. For example, if a brief impulsive noise causes a bit error in a voice application resulting

in packet loss, a user may hear only a “click” or “pop.” However, that same packet loss, under the

same N/I conditions, can completely disrupt a STU-III secure telephone conversation (bit count

integrity is required for the synchronous data stream carrying STU-III voice information).

Alternatively, the Gaussian-type interference caused by a heavily loaded CDMA RF channel may

induce errors that are more uniformly distributed in time, which may be more readily corrected by

higher communication layers. Clearly, the distribution of errors caused by impediments in the PCS

air interface depends on the type of N/I encountered. A comprehensive N/I model is needed.

Complete characterization of the PCS environment includes system-specific models for N/I. As

previously mentioned, the N/I waveform model is comprised of three components: 1) complex,

modulated, zero-mean Gaussian noise; 2) interference from other users of the PCS frequency band;

and 3) impulsive artificial and natural noise. Derivations for the second component, interference

from other users of the PCS frequency band, are presented in this report. Components 1) and 3) are

detailed in [1-6], and are beyond the scope of this report; an informative survey of all three N/I

components in the 2-GHz licensed PCS band is provided in Section 2. A generic method for

wireless cellular interference generation is described in Section 3, and system-specific interference

waveforms for the global system for mobile (GSM)-based PCS 1900, and IS-95-based CDMA are

presented in detail in Sections 4 and 5.

ITS N/I models are tailored for use in real-time channel simulation. To aid evaluation of system

performance, PCS testing includes extensive use of real-time hardware channel simulators. Channel

simulation provides a cost-effective means of testing PCS radio technologies and proposed common

air-interface standards. ITS incorporates hardware channel simulation of the PCS air-interface

propagation environment as part of the PCS testing, modeling, and evaluation program. Future work

will address an efficient hardware implementation of the ITS N/I waveform models.

5

2. NEEDFORNOISEANDINTERFERENCEMODELINGINTHE2-GHZPCSENVIRONMENT

Complete specification of the PCS RF channel must include a credible model of the system-specific

N/I environment. Incidental channel impediments are termed “noise,” and originate from both

natural and artificial sources. ”Interferers,” by contrast, are waveforms intentionally radiated in the

RF channel that disrupt or degrade a desired waveform. Interferers are generated by sources both

external and internal to the affected PCS system, and may include other wireless systems sharing a

common geographical area and frequency band.

Preliminary studies of the noise environment at 2 GHz show that the natural noise contribution is

relatively small. Sources of artificial noise are ranked in order of severity as follows [1]:

1. Automotive ignitions.

2. Transportation and generation facilities.

3. Industrial equipment.

4. Consumer products.

5. Lighting systems.

6. Medical equipment.

Background noise from these sources appears to be very low, even including the automotive ignition

sources. However, point or individual sources may be quite significant noise generators, with noise

levels well above that of background noise. Some indoor environments may also foster noise levels

above background noise. Further investigation is needed in this area [1]. Blackard et al., [2] state that

photocopiers, elevators, and microwave ovens (all elements of the indoor office/retail environment)

are significant sources of impulsive noise in certain PCS frequency bands.

2.1 Interference Environment

Interferers, in contrast to most noise sources, are a major hindrance to wireless communication at 2

GHz. In a multiple access environment using microcellular deployment, the most significant

unwanted waveform sources will be interferers rather than noise sources. For this reason, PCS

systems are interference-limited. PCS system interference is categorized as either external or

internal. External sources of interference are generated by systems outside a given PCS system.

These include microwave links in the PCS frequency band, and other PCS systems sharing the

geographical area. Microwave link interference to PCS systems has been discounted by several

sources [7,8]. However, PCS systems may interfere with microwave links.

7
Intrasystem interferers are (by definition) self-generated by the PCS system, and can be categorized

as either intracellular or intercellular. The nature of both types of internal interference is specific to

a particular technology. For example, time-division multiple access (TDMA) systems do not

experience intracell interference if all users in the cell are properly synchronized. CDMA systems

inherently create intracell interference by overlaying user waveforms within a cell’s frequency

allocation. Furthermore, variations of the different technologies have made the interference highly

system-specific. For example, CDMA systems with joint detection do not experience intracell

interference, at the cost of an SNR degradation [9]. Hybrid CDMA/TDMA systems create their

own unique interference characteristics. Proper modeling of the interferers requires a

system-specific analysis of the waveforms involved.

Development of an N/I model for a hardware implementation requires a quantitative description of

the N/I waveform that must be added to the received waveform. In [6], a general framework for

such a description of the N/I waveform is proposed. The described waveform is represented as the

sum of three components: 1) a complex, zero-mean, modulated Gaussian noise; 2) a summation of

impulsive artificial and natural noise processes; and 3) a summation of interference processes

generated by other users in the PCS frequency band.

2.1.1 Statistical Interference Models

Most descriptions of the interference environment in the literature, e.g., [3, 4, 10, 11, 12], present

statistical models of the interference waveform behavior, rather than a model of the waveform

itself. These models tend to be general to the PCS environment, giving interference statistics in

form generic to any of the many relevant stochastic distributions, in conjunction with desired

waveform distributions, suited to cochannel interference statistics, e.g., Rayleigh, Rician,

lognormal, and Nakagami. Some models [13] describe the total statistical N/I environment, e.g. the

Middleton Class A, B, and C statistical-physical model. Outputs of these models are BER studies,

system outage probabilities, intercell and intracell noise amplitude and phase probability density

functions (PDFs), and others. Models that are technology-specific indicate system performance

based on properties inherent to a particular technology.

Conflicts sometimes exist between these models. One model claims the well-accepted Nakagami

model for the Rician fading of the desired waveform is not appropriate, and the ramifications may

extend into the interference model results [12]. Another model claims that Rayleigh fading may not

be a good assumption for the desired waveform in a microcellular architecture, and thus the fading

statistics of the desired and interferer waveforms must be adapted accordingly [3].

8
A derived statistical model may not be sufficient for complete characterization of the interference

environment. In particular, complex multiple access waveforms from multiple mobile stations may

create an interference environment that cannot be expressed with a simple distribution function. For

example, a Gaussian approximation for cochannel interference from adjacent cells is appropriate

for narrowband TDMA (in certain cases), but may not be valid for a CDMA system with a single

dominant interferer. Accurate modeling of CDMA may therefore involve a complete simulation of

the interference sources [13]. While this may be computationally intensive, models can be

developed that capture the essence of the waveform interference. The simplified waveform then

can be precomputed for use in a real-time hardware simulation.

Power control algorithms also add a layer of complexity to the waveform statistics. For example, a

CDMA system with perfect power control and several concurrent channels in use has an

interference waveform which is approximately Gaussian. However, imperfections in the power

control algorithm may cause a non-Gaussian interference waveform, particularly in shadowing or

“near/far” conditions.

2.1.2 Standards Requirements for Interference Levels in PCS Systems

Many wireless developers use carrier-to-interference power ratios (C/I), or carrier-to-noise plus

interference ratios C/(N+I), to predict the performance of proposed PCS systems. Standards

organizations, such as the Joint Technical Committee (JTC) on Wireless Access, require that the

PCS system developer provide the interference density as a first order statistic, namely the

self-interference power level created by the PCS system. In the JTC, the interference density is

defined as the interference power at a receiver input located at the border of an interior cell.

Interference levels for each proposed PCS system are provided by the system proposer(s), along

with a justification statement [5]. Separate and optional contributions to the JTC suggest

developing C/I as a function of the PCS system load. Adjacent cell interference and external

interference levels are not specified.

Because each system provider specifies the interference level independently, there may be

problems in interpreting the additive effects of intersystem interference. More importantly, because

the interference characteristics of the proposed systems are quantified as a single average power

level, higher-order statistics that may greatly impact system performance are not considered.

Although C/I measures give an indication of PCS system performance in an interference

environment, they may not provide the complete picture.

9
2.1.3 System-specific Models and Interference Simulation

Many systems developers and researchers, faced with limitations imposed by the above

interference measures and models, have turned to simulation techniques for PCS system

performance modeling. Because waveform simulations are computationally intensive, the

simulations are usually software-based and not in real time (references [14-16] give theoretical and

computational analyses of PCS system performance based on Monte Carlo techniques, or system

descriptions that may be adapted to simulation techniques). These analyses tend to be more

system-specific than the statistical models described earlier. Haas, et al. make the assertion that

intercell interference measures have been pessimistic, and that interferers from an adjacent cell may

not cause problems, even with a three-cell reuse pattern [15]. Such claims must be substantiated

through measurement or simulation.

Other studies use system-specific software simulation of the interference sources. In [17], the

TDMA simulation tool BERSIM is used to import channel response data into another simulation

tool that simulates a 1.25-MHz bandwidth CDMA system. Channel processing is accomplished by

downconverting the system waveforms to baseband complex envelope form, convolving the

waveform with the channel impulse response, then adding interference and Gaussian noise. The

discrete nature of the simulation requires sampling of the waveforms involved; in this case, at one

fourth the chip transmission rate. Intracell interferers are assigned a uniformly distributed random

delay and phase, then added to the desired waveform before convolution with the channel impulse

response. All interferers are assumed to have equal amplitudes (a power equalization assumption),

which may be overly simplistic. Intercell interference does not seem to be included in the

simulation. Clearly, a method is needed to develop a sufficiently-detailed interference model that is

computationally viable in simulation.

2.2 Hardware Simulation of PCS Channels in the Noise/Interference Environment

Several manufacturers offer real-time RF channel simulators appropriate to the PCS environment.

These simulators typically have baseband input bandwidths between 5- and 8-MHz, and feature

multipath, propagation loss, and Doppler shift effects. Multipath capabilities usually include at

least six tapped delay line paths. Propagation loss can be chosen independently for each path in

most cases. Some models are developed for particular technologies, such as GSM. In summary,

these simulators have features that characterize the PCS RF channel, with one significant

exception: the N/I environment.

10
Most commercial simulators possess two independent channels that allow cochannel interference

tests via transmission of the interference waveform through a separate multipath channel, and use a

power combiner to add the interferer to the desired waveform. However, these simulators do not

produce the interference waveform itself. (At least one manufacturer provides Gaussian noise

sources for the RF channel, but no interferers are included). Consequently, the interference

waveform must be generated by the PCS system evaluator. This waveform obviously is highly

dependent on the PCS system, but generic models for each technology class can be adapted to

emulate specific PCS systems. For example, models for generic CDMA, TDMA, and hybrid

CDMA/TDMA can be adapted to the many specific PCS systems under consideration in the JTC.

Most likely, PCS system developers model interference waveforms as part of the design process,

but these models tend to be proprietary and nonstandardized between competing technologies.

PCS evaluators need a viable, independent source for the interference waveform model specific to

each system under test. The waveform must be detailed enough to include higher-order statistical

properties that may affect proposed PCS system performance. This may require full software

simulation of the N/I environment, with results that can be adapted to a hardware implementation.

However, the hardware implementation of the interference waveform must also be efficient to

accommodate a real-time interface with a RF channel simulator. Software simulation of the

interference can identify both important waveform aspects, and relatively insignificant elements of

the waveform that add unnecessary complexity to the simulation.

2.3 Summary

System-specific models of N/I waveforms must be developed for PCS tests that use RF channel

hardware simulators. Implementation of the models may include software simulation of the

interference waveform, and adaptation of the results to hardware capable of processing the required

waveforms in real time. Disclosure of PCS system specifications will be required for proper

development of the N/I models. Ideally, the model development and validation will be conducted

by an independent PCS system evaluator with involvement in assessment of each of the PCS

technologies under test.

ITS has developed system-specific interference models for two licensed PCS technologies to date:

the GSM-based PCS 1900, and CDMA-based on IS-95. The method for developing the intrasystem

interference waveforms for these systems, which includes cochannel interference and adjacent

channel interference, is described in the next section.

11

3. PCS CELLULAR GEOMETRY FOR INTRASYSTEM COCHANNEL

INTERFERENCE

Development of an intrasystem interference waveform based on spatial considerations includes

specification of a PCS cellular system geometry. In most analyses of this type, a hexagonal cellular

geometry is used, with each cell having six adjacent cells. Although this geometry is relatively

awkward to manage in simulation, most PCS deployment descriptions use it. For ITS models, the

single base station is located at the center of the cell, but a circular geometric pattern is assumed. As

shown in Figures 3.1-3.7, cells have been augmented to be circular. This results in some minor

overlap, but makes the geometry easier to manipulate. This representation actually reflects reality

since base stations in “real” cells will not radiate hexagonal patterns, and some overlap is expected

for complete area coverage. Overlap also allows future addition of noncircular cell patterns, such as

those created by directional antennas.

3.1 Assumptions

The following assumptions are made for the PCS cellular geometry: 1) all cells within the PCS system

are the same size, 2) each cell contains Nu mobile stations, 3) one RF carrier is simulated per cell, and

4) a primary cell contains the base station or mobile stations experiencing the interference.

The primary cell is the focus of the interference waveform simulation. Because the licensed PCS

technologies use frequency division duplexing (FDD), interference waveforms must be developed

for both the uplink and downlink. On the uplink, it contains the base station experiencing

interference from mobile stations outside the primary cell transmitting on the same RF. In CDMA

systems, mobile stations within the primary cell also contribute to the interference waveform. The

primary cell also contains mobile stations that receive interference from nearby base stations

transmitting on the same downlink RF carrier as the primary cell base station. Uplink and downlink

cases for the intrasystem cochannel interference waveform are summarized as follows:

1. Uplink interference (two components)

Component 1: Mobile stations in interfering cells create an interference waveform at

the primary cell’s base station. This is the intercell interference component.

Component 2: Mobile stations within the primary cell interfere with each other via

intracell interference. This is an inherent property of CDMA systems; TDMA systems

with timing problems may also demonstrate intracell interference.

2. Downlink interference (conditionally two components):

13
Component 1: Base stations in interfering cells create interference at each of the

mobile stations in the primary cell, forming the intercell interference component of the

interference waveform.

Component 2 (conditional): The base station in the primary cell generates a multiplexed

waveform that interferes with other mobile stations in the primary cell. This should only

be a problem in CDMA systems; TDMA waveforms will be internally combined within

the base station, so synchronization of time slots should not be a problem.

Component 2 in both cases above may have to be generated by patching real-time outputs of the

actual PCS transmitters into a channel simulator, with proper multipath, delay, Doppler shift, and

propagation loss factors. This is readily apparent for TDMA systems, since (for uplink transmissions)

a mobile station under test must synchronize itself into a time slot assigned by the base station. This

requires the base station to have control of time slot assignments for the entire uplink (i.e. all control

channels which are dynamically allocating channels to the mobile stations). This cannot be simulated

by blindly filling the other time slots with modulated energy. Prerecording actual mobile station

waveforms for insertion into the channel simulator may prove difficult, if not impossible because of

the state-dependent nature of the PCS medium access protocols.

Figure 3.1 shows the geometry for a confederation of cells in the PCS system. It is independent of

any specific PCS system technology. Each cell is modeled as a circle with the base station at the

geometric center, with the familiar cellular hexagon subscribing the circle. RC is defined as the

radius of the base station circle, and RH is the “short radius” of the hexagon subscribing the base

station circle. Relationships between RC and RH are shown in Figure 3.2.

3.2 Interference Waveform Notation

Ferranto and Lemmon describe a method for representing an interference waveform from a nearby

cell based on Cartesian coordinate translation of the desired waveform expression [18]. This

description is used in the following discussion, with some minor notational changes.

The primary cell, as defined in Section 3.1, is surrounded by 6 adjacent interfering cells, and 12

additional interfering cells surround the primary cell and 6 adjacent interferers. The distance from

and angle to the n

th mobile station relative to the primary cell’s base station are defined as rp n, and

 p n, , respectively. N U mobile stations are present in the primary cell. x p n, and y p n, are the Cartesian

x and y distances, respectively, of the n

th mobile station from the base station. These relationships

are shown graphically in Figure 3.3.

14
15

Figure 3.1. Hexagonal cells in a PCS deployment with circular geometry overlay.

RH

RC

RC

R /2 C

30°

R = radius of circle

R = “short radius” of hexagon = R cos 30 C

H C °

Figure 3.2. Geometry and notation for a single PCS cell.
As a framework for the system-specific interference waveform expressions, generic waveform

terminology is derived in terms of complex baseband representations of the interference. Complex

baseband notation used in the remainder of this report is described in detail in Appendix B. Waveforms

in complex baseband form are denoted by an overline. Expressions are defined as follows:

x t FI k n ( ,) ( )is the complex baseband expression for the interference waveform from the n

th interferer

in interfering cell k (“F” indicates that the expression is defined over one “frame” for the downlink

case, and the “I” represents the “interferer”). Depending on the context, the interferer may be either

a mobile station, or a base station. Two cases exist:

1. For uplink, n represents the n

th interfering mobile station in the kth cell. For TDMA

systems, x t FI k n ( ,) ( ) is defined over one time slot. For CDMA systems, x t FI k n ( ,) ( ) is not

time-limited.

2. For downlink, n indicates the n

th mobile station in the primary cell experiencing

interference from the base station in interfering cell k. x t FI k n ( ,) ( ) is defined over one

frame for TDMA systems, and is not time-limited for CDMA systems.

16

p,n

MSNU-1

rp,n

xp,n

yp,n

MS0

MS1

BS

MSn

Figure 3.3. Geometric relationships between base stations and mobile stations within a single

PCS cell.
x t FI k n ( ,) ( )is the basic building block for the interference waveform expressions. In the N/I model, it

is the only expression referenced to the transmitter (i.e., all other expressions represent the

interference waveform incident at the receiver). All other interference functions are expressed as

summations of x t FI k n ( ,) ( ). For example:

x t FI k(,) ( ) is defined as the sum of all uplink interferer mobile station waveforms created in

interfering cell k, incident at the primary cell’s base station.

x t FI ( ) is the aggregate uplink interference waveform incident at the primary cell’s base station.

x t FI n ( ,) ( ) is the downlink interference waveform caused by all interfering base stations, as

seen by the n

th mobile station in the primary cell.

Expressions for the uplink and downlink interference functions depend on the technology.

However, generic expressions can be formed with parameter specification of a CDMA- or

TDMA-based technology. These generic expressions are shown below, with the parameter b

specifying CDMA or TDMA:

For uplink,

x t x t nbT

r

c

a r FI k FI k n U

k n

( , ) ( ,) kn kn k

,

,, , () ,  



 



	 
   n

n

N

FI FI k

k

N

U

I

xt x t













 

0

1

0

1

( ) ( ), (,)

(3.1)

and for downlink,

xt xt 
  r

c

a r FI n FI k n

k n

kn kn kn

k

N

( ,) ( ,)

,

,, , () , 



 



 



	 

0

I 



1

, (3.2)

where, for both Equation (3.1) and (3.2):

c is the speed of radio wave propagation,

TU is the duration of a TDMA time slot,

N U is the number of mobile stations (users) per cell,

N I is the number of interfering cells,

rk n, is the distance between the point interferer and the interference reception point (this

is defined in detail in the next section),

k n, is the angle of the point interferer with respect to the interference reception point

(this is defined in detail in the next section),

a r kn kn kn ,, , (, )  is a propagation loss and Doppler shift function,

r

c

k n, is the propagation delay, and

b indicates TDMA (b = 1) or CDMA (b = 0).

17
The parameter b allows flexibility in the notation to accommodate both TDMA and CDMA. For

TDMA systems, each communication link between mobile station and base station makes full use

of the RF channel during its assigned time slot; b then “activates” the time delay in x t FI k(,) ( ) . For

CDMA systems, each communication link shares the same time and frequency domains, and

therefore communication links overlay one another.

Note that x t FI k n ( ,) ( ) must be defined within the specifications of a particular PCS technology. For

example, the single interferer represented by x t FI k n ( ,) ( ) is defined over one time slot in Equation

(3.1) for TDMA, but is defined over one frame in Equation (3.2) for TDMA. For CDMA, frames

and time slots (in the context of interference generation) become meaningless, and x t FI k n ( ,) ( ) is

defined over an arbitrary period of time. For example, an IS-95-based CDMA system may define

x t FI k n ( ,) ( ) in terms of chip duration or spreading code length.

3.3 Intercell Uplink Interference

By performing translations with respect to reference Cartesian coordinate systems, expressions for

interference to the primary cell from nearby cells can be derived. Interference waveforms all have

the same form as the desired waveforms derived for the primary cell, with the exception of the

coordinate translation, because the interferers are assumed to belong to a common PCS system.

This of course assumes the nearby cells are using the same modulation scheme and system

configuration as the primary cell. However, the aggregate interference expression may be imposed

upon any system under test. For example, an interference waveform created by a PCS 1900 system

may be applied to an IS-95-based PCS system (or any system) under test.

3.3.1 Uplink Cell Geometry

Uplink interference is caused by mobile stations in nearby cells (and in the primary cell for CDMA)

transmitting unwanted waveforms to the base station in the primary cell. The k

th interfering cell is

defined graphically in Figure 3.4.

In Figure 3.4,

d dd k x y k k  
 2 2 is the distance between the primary cell’s base station and the k

th interfering

cell’s base station,

x y kn kn , , , are the x and y distances of the n

th mobile station in the kth interfering cell from

the k

th cell’s base station, respectively,

18
k is the angle of interfering cell k’s base station with respect to the primary cell’s

base station,

k n, is the angle of the n

th mobile station in interfering cell k with respect to the

primary cell’s base station, and

rk n, is the distance of the n

th mobile station in interfering cell k with respect to the

primary cell’s base station.

From simple geometry:

r xd yd kn kn x k n y k k ,, ,  


 ( ) ( ), 2 2 (3.3)

and

k n

k n y

k n x

y d

x d

k

k

,

,

,

arctan ( )

( )  . 













 (3.4)

19

dk

Primary cell

k interfering cell th

k

BSP

k,n

BSk

dxk

dyk

rk,n

xk,n

yk,n

MSk,n

Figure 3.4. Cartesian coordinate translation for an interfering cell.
3.3.2 Adjacent Cell Distances

In the hexagonal pattern shown in Figure 3.1, cells adjacent to the primary cell are all equidistant, and

the primary cell’s base station is separated from each adjacent cell’s base station by distance dk ,

where

dR R k kHC   



 

 2 2 	  

6 cos , . 0 5  (3.5)

Plotting the adjacent cell base station locations on the Cartesian coordinate system with the primary

cell’s base station at the origin yields the geometry shown in Figure 3.5.

Clearly, in Figure 3.5,

 

k   k k

3

, . 0 5 (3.6)

So for adjacent cells only,

dd R k k

d d

x kkC

y k

k

k

  



 



	 



 



	  



cos cos cos ,    2

6 3

0 5

sin cos sin , .   

k C R k  k 



 



	 



 

 2 	  

6 3

0 5

(3.7)

20

BS0

d , 0 0 

x

d , 1 1 d , 2 2  

d , 3 3 

d , 4 4  d , 5 5 

BS BS 1 2

BS3

BS4 BS5

y

Figure 3.5. Geometric relations of adjacent cells.
Equation (3.7) can then be substituted into Equations (3.1), (3.3), and (3.4) to find the uplink

interference waveform from adjacent cells. In Equation (3.1), the number of interfering cells NI is

set equal to six.

3.3.3 Second-level Cell Distances

The second layer of interfering cells from the primary cell are not equidistant, as was the case for

cells adjacent to the primary cell. Of the twelve cells that comprise the second layer, half are at

distance 3RC (as measured from the primary cell base station to an interfering cell base station), and

half are at distance 4 4 R R H C 6  cos , where RC and RH are as defined in Section 3.1. Numbering for

the second-layer cells is arbitrary. Numbers 6 through 17 are used to distinguish second-layer cells

from adjacent cells. Figure 3.6 demonstrates the relationships graphically. In Figure 3.6, for even k:

d R

k

d R

d

k

E

C

k

x

E

C k

y

E

k

k



 



4

6

6

4

6

6

cos

( ) ,

cos cos





 



where

 4RC k 6 cos sin ,  

(3.8)

and for odd k,

21

BS6

d , 6 6 

x

d , 8 8 

d , 7 7 

d , 17 17 

BS8

BS7

BS17

y

...

...

Figure 3.6. Second-layer interfering cell distances.
d R

k

d R

d R

k

O

C

k

x

O

C k

y

O

C k

k

k



 





3

6

6

3

3

 





( ) ,

cos

sin .

where

(3.9)

As in Section 3.3.2, Equations (3.8) and (3.9) can be substituted into Equations (3.1), (3.3), and

(3.4) to find the interference waveform from the second layer of cells surrounding the primary cell.

In this case, NI will equal 12 in Equation (3.1). The total uplink interference waveform is the sum of

the adjacent and secondary interference waveforms. This analysis can be extended to the third,

fourth, and greater layers of surrounding interfering cells, but doing so makes the interference

waveform computation much more intensive. Layers beyond the second layer are typically far

enough away to be discounted in the interference computation. For this reason, ITS analysis

employs adjacent interfering cells and second-layer interfering cells only.

3.4 Intercell Downlink Interference

Analysis for intercell downlink interference is similar to the analysis for intercell uplink

interference as described in Section 3.3. Intercell downlink interference is caused by base stations

in nearby cells interfering with mobile stations in the primary cell. For CDMA systems, the base

station in the primary cell may cause additional interference. In this section, the intercell downlink

interference expressions are described.

3.4.1 Downlink Cell Geometry

Intercell downlink interference geometry is quite similar to the geometry specified for intercell

uplink interference; in fact, the geometries are complementary. The k

th interfering cell geometry is

shown in Figure 3.7.

In Figure 3.7,

d dd k x y k k  
 2 2 is the distance between the primary cell’s base station and the k

th interfering

cell’s base station,

x y pn pn , , , are the x and y distances of the n

th mobile station in the primary cell with respect

to the primary cell’s base station,

k is the angle of interfering cell k’s base station with respect to the primary cell’s

base station,

22
k n, is the angle of the base station in interfering cell k with respect to the n

th mobile

station in the primary cell, and

rk n, is the distance of the base station in interfering cell k with respect to the n

th

mobile station in the primary cell.

From simple geometry,

r dx d y k n x pn y pn k k , , ,  
( ) ( ), 2 2 (3.10)

and

k n

y pn

x pn

d y

d x

k

k

,

,

,

arctan ( )

( )  . 











 (3.11)

3.4.2 Adjacent Cell Distances

Adjacent cell distance calculations for the downlink interference waveform are identical to those

detailed in Section 3.3.2. The downlink interference waveform component from adjacent cells can

be found by substituting Equations (3.5), (3.6), and (3.7) into Equations (3.2), (3.10), and (3.11).

23

Primary cell

k interfering cell th

dk

BSP

BSk

rk,n

k

k,n

xp,n

yp,n

MSk,n

dyk

dxk

Figure 3.7. Cartesian coordinate translation for an interfering cell.
3.4.3 Second-level Cell Distances

Second-level cell distance calculations for the downlink interference waveform are identical to

those detailed in Section 3.3.3. The downlink interference waveform component from second-level

cells can be found by substituting Equations (3.8) and (3.9) into Equations (3.2), (3.10), and (3.11).

3.5 Summary

In this section, generic expressions for intrasystem interference were developed using a cellular

geometry. In Sections 4 and 5, system-specific interference waveforms based on these generic

expressions are presented for GSM-based PCS 1900, and IS-95-based CDMA PCS.

24
4. GLOBAL SYSTEM FOR MOBILE-BASED PCS 1900 INTERFERENCE

WAVEFORM

4.1 PCS 1900 Physical-layer Overview

PCS 1900 is a narrowband TDMA system based on GSM, and designed for use in the 2-GHz

licensed PCS frequency band. It offers a range of digital services for public applications, including

both voice and data. JTC contribution [19] specifies the air-interface standard for PCS 1900, and

was used for all PCS 1900 interference model derivations in this report.

PCS 1900 is a complete mobile system. As such, its standard includes all layers of communication,

including networking, interworking with fixed networks, and other higher-layer signaling

protocols that have indirect impacts on the interference model. Although the ITS interference

model for PCS 1900 uses the physical-layer specification exclusively, these indirect impacts also

have an effect on the interference environment. A certain deployment scenario, as managed by

higher-layer entities, may result in increased channel use and therefore increased system

interference. These issues are handled by the link- and network-level simulation as part of ITS

methodologies described in Section 1. The physical-layer interference model is developed with

these higher-layer issues in mind.

To obtain a PCS 1900 interference waveform, a model of the physical layer, as specified in [19], is

developed. PCS 1900 employs 200-kHz RF channels in a frequency division duplexing scheme.

The licensed uplink band (1850-1910 MHz) and downlink band (1930-1990 MHz) may each

theoretically support up to 298 PCS 1900 RF channels (not including two 200-kHz guard bands),

although frequency reuse, frequency hopping algorithms, and specific equipment designs restrict

the number of concurrent active RF channels in a given cell. PCS 1900 TDMA is based on eight

time slots per frame, with 156.25 bit durations allocated per time slot, including guard times. Each

time slot lasts 0.5769 ms, therefore the total modulation rate is approximately 270.8 kb/s. Each

transmission within an assigned time slot is called a burst. The data rate per user depends on the

type of burst used within each time slot, and the number of time slots used per frame per user.

Figure 4.1 shows the reference configuration for the PCS 1900 physical layer. This includes

channel coding, interleaving, and encryption, which do not directly contribute to the interference

waveform. For noise and interference modeling purposes, the data content of the interferer is not

important, as long as sufficient randomness exists in the data stream. Important factors are the

modulation scheme, transmitted power levels, propagation environment, and geometric

distributions of the interferers. Interference modeling techniques allow abstraction of all

25
communication devices that manipulate the data stream before modulation. Abstracted

communication devices are contained within the dotted line in Figure 4.1. These form the

“transmission data” block shown in Figure 4.2.

4.2 Theoretical Definition for PCS 1900-based Gaussian Minimum-Shift Keying

Gaussian minimum-shift keying (GMSK) is used for PCS 1900 base station and mobile station

transmitters. Theoretical definitions for PCS 1900-based GMSK are presented in [19], and are

defined in terms of the N/I waveform notation presented in Section 3. These definitions do not

allow practical waveform generation as specified, but their development is necessary before

computer simulation and hardware implementation for a channel simulator is possible. The

practical implementation is described in Sections 4.6 and 4.7.

Transmission data output sequences are differentially encoded via modulo two addition given by

26

Information

bits in Block

Encoder

Convolutional

Encoder

Reorder

and partition

Interleaver Encryption Burst

Multiplexing

Differential

Encoder

GMSK

Modulator Transmitter

To air

interface

Figure 4.1. Physical-layer configuration for PCS 1900 (mobile station or base station).

Information

bits in Transmission

Data

GMSK

Modulator Transmitter

To air

interface

Figure 4.2. Abstracted physical-layer configuration for PCS 1900.
 ii i   d d 1 , (4.1)

where di [,] 01 represents the i

th data bit value. The modulated bipolar value then becomes

  i i  1 2 , where  i  [ ,] 11 , which is passed to the GMSK modulator.

Notation used for a generalized phase-modulated waveform is developed in Appendix A. The

GMSK phase term is given by:



 ( , ) ( ), t q t iT  i

i

  



2 (4.2)

where the modulation index h is equal to 0.25 in Equation (A.4).3 The pulse q t( ) is expressed as

follows:

q t g u du

u

t

() ( ) , 

 



 



	



 where

g(t) = h(t) rect t

T

h(t) = 1

2

and

= ln(2)

2 BT

 





 T

t

T exp ,

.

 











2

2 2

(4.3)

In Equation (4.3), modulating bit rate1 / T is 270.833 kb/s, and the time-bandwidth product TB is 0.3.

In addition, a normalization constraint requires that q t( ) converge to unity as t approaches infinity.

The theoretical modulated RF carrier waveform for PCS 1900-based GMSK is converted to

complex baseband form (see Appendix B) for implementation in the noise and interference model.

The GMSK waveform output from the transmitter is expressed in bandpass form as:

x t ! E

T ft t s

c ( , ) cos ( , ) ,    



2 2 0 (4.4)

where (, ) t  is defined in Equation (4.2), f c is the carrier frequency, Es is the energy/modulating

bit, and 0 is a random phase term that is constant over the entire mobile station’s assigned time

slot. Using notation developed in Appendix B,

27

3 In [19], the modulation index h is specified as 0.5. The scale factor is different because the

phase expression in Equation (4.2) is defined with a multiplier of 2, instead of the scale

factor defined in [19].
 !

 !

 !

xt x t

xte

Rte e

j ft

j t j f

c

c

( ) Re ( )

Re ( )

Re ( ) (, )













2

0 2



 

 !t .

(4.5)

Therefore R t E

T

s ( )  2 , and

x t ! E

T jt j s ( , ) exp ( , ) ,    


2

 0 (4.6)

which represents the PCS 1900-based GMSK complex envelope from a single transmitter. This

expression applies to both base station and mobile station transmitters.

4.3 PCS 1900 Modulated Waveform Generation

One approach to PCS 1900-based GMSK waveform simulation is to develop the pulse  i

g t 
  for

each possible input symbol. These pregenerated pulses then can be transmitted in a random order, as

specified by the output data stream from the abstracted transmission data block shown in Figure 4.2.

For GMSK, the two possible pulses, corresponding to  i  ! 11, , are generated by exciting a

pulse-shaping filter with impulse response g t( ). Excitations corresponding to the two possible

symbol values are represented by ut t 1 () ()  " and ut t 
1 () ()  " , where "( )t is the Dirac delta

function. For ut u t t () () ()   
1 " ,  
1 gt gt () ()  . Similarly, for ut u t t () () ()  1 " ,

 1 gt gt () ()  . The pulse-shaping filter is shown graphically in Figure 4.3.

Pulses equal to #g t( ) are precomputed using Equation (4.3), and transmitted every T seconds,

where g t( ) represents input symbol  i  1, and g t( ) represents input symbol  i  1. The pulses

g t( ) are used to modulate the instantaneous frequency f t I ( ) given by Equation (A.5). In an actual

hardware implementation, this may be accomplished with an FM modulator, or by using lookup

table techniques. For simulation, (, ) t  is represented mathematically by Equations (4.2) and

(4.3).

28

u(t) g(t) PCS1900 GMSK

pulse-shaping

filter, g(t)

Figure 4.3. PCS 1900 GMSK pulse generation.
Everything prior to the generation of (, ) t  in Equation (4.2) can be precomputed. The following

section presents time-domain expressions for ( , )| t  i0 corresponding to each of the two possible

symbol values. These then may be implemented as a component of the complex baseband

expression for the GMSK waveform by appropriate delaying and summing of individual phase

terms. This is demonstrated in Section 4.7, which details the simulation methodology.

4.3.1 Time-domain Representation of the Phase Expression

The PCS 1900-GMSK phase is most easily expressed in the time domain. Again, the PCS

1900-GMSK pulse g t( ) defined in Section 4.3 is the convolution of the Gaussian filter h t( )

described by Equation (4.3) with the rectangular function, gt ht 
 t

T () ()   rect . The convolution

can be accomplished by remaining in the time domain, then solving the resulting integral. The

convolution may also be solved as a multiplication in the frequency domain, but a nontrivial inverse

Fourier transform of the phase term makes this impractical. Expressing gt ht 
 t

T () ()   rect in

Equation (4.3) as a convolution integral,

gt h t

T () ( ) .  d  



 



	 



 $

$

$

$

rect (4.7)

To eliminate the rect[·] function, note that rect
  t $

% equals 1

T over the range & & 1 2

1

2

t $

% , or

t t T T & &
 2 2 $ . Then g t( ) becomes:

g t Th d

t T

t T

() ( ) . 

 




 1

2

2

$ $

$

(4.8)

Substituting Equation (4.3) into Equation (4.8), and defining constants a and b for Equation (4.3) as

follows,

ht b t

a

a T

b

T

( ) exp ,

,

,

' 



 



	





2

2

2 22 2

1

2

where

 and



(4.9)

g t( ) is then expressed as

g t

b

T a

d

t T

t T

( ) exp .  



 



	

 




 $

$

$

2

2

2

2

(4.10)

Changing the integration variables to s a  $ and ds d

a  $ , Equation (4.10) becomes,

g t 
  ab

T

s ds

s t

a

T

a

t

a

T

a

( ) exp .  

 




 2

2

2

(4.11)

29
Next, g(t) is expressed as two integrals by splitting the region of integration,

g t 
  
  ab

T

s ds ab

T

s ds

s t

a

T

a s t

a

T

a

( ) exp exp .  

 




 



  2 2

2 2

(4.12)

The integrals in Equation (4.12) can be expressed in terms of the complementary error function,

which is defined as:

erfc( ) exp . w s ds 
  s w 





 2 2



(4.13)

Substituting Equation (4.13) into Equation (4.12), and replacing the constants a and b defined in

Equation (4.9), the expression for g t( ) becomes:


  
  g t T

Bt Bt T T

( ) ln ln   











	

	  




 1

2

2

2 2

2

2 2

2 2 erfc erfc

 







	

	











, (4.14)

where TB is defined in Equation (4.3). For i  0, the PCS 1900-GMSK phase term is expressed as:





$ $

$

( , )| ( ) . t gd i

t

 



0  2 (4.15)

Values other than i  0 may be simulated by shifting the phase expression by T seconds for each

symbol. The aggregate phase expression is computed by summing Equation (4.15) over all values ofi.

Note that the integration in Equation (4.15) cannot be solved analytically. It must be computed

numerically, or solved by expansion in a power series. The ITS N/I model uses numerical

integration to precompute the phase for each of the two possible symbols. The phase component

created by each symbol must be time-limited for computer simulation. In a real PCS

implementation, the Gaussian pulse g t( )will be truncated symmetrically to an even integer multiple

of length T. For the ITS N/I model, the Gaussian pulse is truncated such that it is nonzero only on

the interval ! && 2 2 Tt T . Consecutive symbols are simulated by shifting the pulse by T seconds

for each symbol after precomputation, then summing over all phase components.

As an option, the phase expression may be approximated by a power series. Although the ITS

model does not employ the power series approximation, the derivation is presented as an alternative

time-domain implementation. This method is presented for those who wish to avoid a numerical

integration. However, the final expression (given in Equation (4.23)) proves to be much more

difficult to implement than the numerical integration method described in Section 4.3.2.

30
First, the complementary error function expression for g t( ) in Equation (4.14) is converted to the

error function. Since erfc erf () () w w  1 , Equation (4.14) can be written as:


  
  g t T

Bt Bt T T

( ) ln ln   












	

	  







1 

2

2

2 2

2

2 2

2 2 erf erf

 



	

	











, (4.16)

and letting


  
  u = erf and u = erf 1 2

2

2 2

2

2 2

2 2   Bt Bt T T  












	

	

 









ln ln 

	

	

, (4.17)

where it is understood that u1 and u2 are both time functions, Equation (4.16) becomes:

g t ! T () ( ) ( ). ' u u

1

2 1 2 erf erf (4.18)

Erf(x) can be expanded into a power series, given by [20],

erf( ) ( )

( )!( )

x , x

k k

k k

k

 




 





2 1

12 1

121

 1

(4.19)

which is integrable. Then, substituting Equation (4.18) into Equation (4.15) yields:

 ! $

$

( , )| ( ) ( ) . t

T

u ud i

t

 



 0 12  4

erf erf (4.20)

Combining Equations (4.19) and (4.20) results in:



 ( , )| ( )

( )!( )

t

T

u

k k i

k k

k

 


 





 

0  

1

1

2 1

1 1 4 1

1

12 1

1 1

1

( )

( )!( ) . 
















 







  1

12 1

2 2

2

1

2

2 1

2 2 1

k k

k

t

u

k k

d$

$

(4.21)

By letting nT be the lower limit of the integration, where n is the integral number of modulated

symbol durations T on each side of the origin for the 0th symbol only, Equation (4.21) becomes

(substituting for u1 and u2 ):

31

 





 $

( , )|

( ) ln

(

t

T

B

k i

k

T k

 








 












	

	

0

1 2

2 1

1 2

1

2

2 2

1

1


 

12 1

2

1

2

2 2

1 1

1 2

1

2

)!( )

( ) ln

k

d

T

B

k nT

t

k

T



 












  $



 $

$









	

	











 

2 1

2 2 1

2

2 12 1

k

k nT

t

k k

d

( )!( )

$.

$

(4.22)

which may then be integrated as follows:


 





 $

( , )| ln

( ) ln

t

BT

B

i

k

T k

 






 












	

	

0

1 2

2

2 2

4

1

2

2 2

2

1

1


 

kk k

B

k

k

T

11 1 1

1 2

12 1

1

2

2 2

1

2

( )!( )

( ) ln



 











	

	










 $

2

22 2 1

2

2 2 12 1

k

k n kk k ( )!( ) 







































 





 $ T

t

(4.23)

4.3.2 Calculation of the PCS 1900-GMSK Phase Pulse Using Numerical Integration

In the previous sections, expressions for the phase of the PCS 1900 GMSK waveform were derived

in terms of the phase pulse g t( ) in the time domain. Theoretically, g t( ) is defined over the interval

 !  & &  t , and g t( ) must also be integrated from negative infinity to t to obtain the phase

component from each transmitted symbol in every PCS 1900 time slot. Of course, this is impossible

to implement in a channel simulation directly; therefore, approximations that reflect actual PCS

1900 modulator and transmitter operation must be made.

The noise and interference model is precomputed g t( ) and the integration is performed numerically

in Unix-based software using the time-domain expressions described in Section 4.3.1. This is not as

time-consuming as it sounds, because only two possible symbol values exist for the PCS 1900

GMSK system, corresponding to 
g t( ) and g t( ). This results in a one-time single integration

accomplished separately from simulations that use the phase expression. Other higher-level

simulations use time-shifted versions of this precomputed phase expression.

Note that the precomputed integration of g t( ) corresponds to the phase component resulting from a

single PCS 1900-GMSK symbol. Because GMSK modulation possesses memory properties, the

actual phase value for each transmitted symbol duration will be the accumulated sum of phase

values from previously transmitted symbols within the PCS 1900 time slot. The resultant phase

32
value is computed by higher-level simulation as the sum of all precomputed and time-shifted phase

components, and is not part of the precomputation routine.

To precompute the phase expression component from a single PCS 1900-GMSK transmitted symbol,

several pieces of information are needed. These include the modulated symbol duration, the 3-dB

bandwidth of the Gaussian filter G f ( ) corresponding to impulse response g t( ), the integer number n

of modulated bit durations needed to integrate g t( ), and the number of samplesN needed to accurately

represent g t( ). Equation (4.18) is the starting point, which is repeated here for convenience:

g t ! T ( ) ( ( )) ( ( )) .  ut ut

1

2 1 2 erf erf (4.24)

Functions u1 and u2 are given in Equation (4.17). To obtain    ( , )| ( , ) ( ) t tt  i0 0  (

, which

represents the phase value due to the i th  0 symbol value only, g t( )must be integrated numerically

over a finite interval.

Practical GMSK modulators integrate the Gaussian pulse over an integral number of modulated

symbol lengths. This is accomplished in the simulation by phase precomputation. Phase

precomputation is realized by iteratively integrating g t( ) from nT to t, increasing t for each

iteration to approximate a continuous time function. If T is the symbol duration, the phase

expression due to the i th  0 symbol becomes (using Equation (4.20)):

 ! $ $$

$

( ) ( ( )) ( ( )) . t

T

u ud

nT

t

 



 4 1 2 erf erf (4.25)

Substituting u t 1 ( ) and u t 2 ( ) in Equation (4.25),

u u

B T

u u

B T

1 1

2 2

2

2 2 2

2

2 2 2

( ) ln

( ) ln

$



$

$



$

 
 



 



	

 



 



	

(4.26)

and

du B d du 1 2

2

2 2   

$

ln (4.27)

yields

33

 





 

( ) ln ( )

ln

t

T B erf u du

u B nT T

 











	

	

 


4

2 2

2 1 1

2

2 2

2

1 2


  







B t

u B

T

T B erf u du

2 2

2 2

2

2 2

2

2

4

2 2

2

ln

ln

ln ( )






 











	

	


 


 







nT

B t

T

T

2

2

2

2 2



ln

. (4.28)

A computer program was written to iteratively perform the integration in Equation (4.28) from nT

to t mT  p , where m is an iteration counter defined from m  1 to m N , and Tp is the phase

expression’s sampling interval. An adaptive-recursive Newton-Cotes algorithm was used for the

integration. Figure 4.4 shows the phase pulse g(t); Figure 4.5 displays the resulting phase after

performing the integration in Equation (4.28).

4.4 Power Considerations

In addition to geographic distribution and modulation type, the output power level of PCS 1900

transmitters must be quantified in the N/I model. The output power of a PCS 1900 mobile station or

base station transmitter is defined as the time average over the useful part of a burst (time slot), as

measured at the output to the passive radiation apparatus. This does not include dummy bits

transmitted during ramp-up and ramp-down transitions at the beginning and at the end of a burst,

34

Figure 4.4 GMSK phase pulse, g t( ).
respectively. Calculations and expressions below reflect the power levels at a transmitter reference

point. This reference point is used for all output waveform amplitude calculations in the ITS N/I

model. Figure 4.6 shows the power measurement point X at the transmitter.

The average power per PCS 1900 burst is defined by:

P ! T

E

T f t t dt ET av B

B

S

c

t

T

s

B

,  
 cos ( , ) ,



 1 2 2 2

0

   (4.29)

where TB is the time duration of the burst, ES is the energy per symbol, andT is the symbol duration.

As expected, the constant envelope expression yields a constant average power value.

For the mobile station, the maximum transmitted power,
  E

T

S

max , is 2W, as specified in [19]. The

PCS 1900-based GMSK complex envelope defined in Equation (4.6) then becomes, for a

maximum power transmission from a single mobile station:

35

Figure 4.5 Resulting phase p t( ) after integration of Equation (4.28).

BS or MS

Transmitter

Active

Antenna

Elements

X Passive

Antenna

Elements

Figure 4.6. PCS 1900 reference diagram showing the power measurement point, X.
xt j t j ( , )| exp ( , ) .   max,MS  
 2 !  0 V (4.30)

Likewise, the maximum transmitted power from a PCS 1900 base station per RF carrier
  E

T

S

max

, is

equal to 40 W, with the total power output from all RF carriers transmitted from the base station not

exceeding 1000 W. The PCS 1900-based GMSK complex envelope defined in Equation (4.6) for a

maximum power transmission on one RF carrier from a single base station is then given by:

xt j t j ( , )| exp ( , ) .   max,BS  
 80 !  0 V (4.31)

Maximum power requirements provide an upper bound for the N/I waveform. The actual

interference power incident at the receiver under study will typically be the sum of multiple

interferers, with each interferer subject to channel impairments via a separate propagation path.

Channel impairments and losses, which are functions of distance, location, and velocity, are

included in the N/I model after determining output waveform levels at the power reference point.

4.5 Timing and Synchronization

For a receiver operating in a multiple access wireless environment, waveform acquisition involves

some method of gaining timing and synchronization before information is transmitted on the link.

TDMA systems, such as PCS 1900, will maintain symbol, slot, and frame synchronization on all

logical channels on the forward link because the waveform is combined within the base station

before transmission. Symbol synchronization is not assumed between different base station

transmissions on the forward link.

However, the reverse link multiple access waveform symbols are generally not synchronized from

time slot to time slot. Each mobile station will transmit its modulated symbol stream with a random

phase term that is constant over the entire time slot. In addition, geographic distribution of mobile

stations within the cell will result in a time offset of the symbol stream that is proportional to the

distance from the base station. For these reasons, a randomU , 
 ! 0 2 phase term andU T 
 ! 0, offset is

added to each symbol stream on the reverse link.

4.6 Interference Expressions

In Section 3, the PCS cellular geometry for intrasystem interference was described, and

technology-independent interference waveform expressions were developed. In Section 4 up to this

point, a physical-layer description of a single PCS 1900 modulator was given. For characterization of

the total intrasystem N/I environment for PCS 1900, the description of the single PCS 1900 modulator

36
must be extended to represent the numerous interferers typically encountered at any given time during

a call session. This is accomplished by duplicating the single PCS 1900 modulator expression once

for each of the interferers, and distributing them via the geometric expressions described in Section 3.

In addition to geometric distribution, the expressions of Section 3 also incorporate delay, propagation

loss, and Doppler shift functions that may be chosen arbitrarily. Multipath may be included by

duplicating each single interferer N p times, where N p is the integer number of paths. Each path must

be delayed and weighted as specified by an appropriate multipath model.

Two cases are considered for the PCS 1900-specific interference waveform: 1) uplink intercell

interference, and 2) downlink intercell interference. Because PCS 1900-GMSK modulation is used

for both base station and mobile station transmissions, the modulation expressions described earlier

in Section 4 apply to both uplink and downlink; only the transmitted power level differs.

4.6.1 Uplink Interference Expressions

Uplink intrasystem interference from PCS 1900 is caused by mobile stations in nearby cells

radiating unwanted energy into the receiver frequency band of the primary cell’s base station. PCS

1900 generates a waveform with eight time slots per simplex RF carrier, hence the maximum

number of interfering mobile stations is eight/cell/RF carrier. Adjacent RF channels may be

considered by frequency-shifting the entire interference waveform by the adjacent channel

distance, which is 200 kHz for PCS 1900. Mobile stations in the primary cell all should be

synchronized to the proper assigned time slot.

Equation (3.1) provides generic expressions for uplink intrasystem interference. Because PCS 1900

is a TDMA technology, the parameter b is given the value 1 to simulate the time slot offsets from

each mobile station in an interfering cell. Equation (3.1) then becomes:

x t x t nT 
  r

c

a r FI k FI k n U

k n

( , ) ( ,) kn kn kn

,

,, , () ,  



 



	 


 

n

N

FI FI k

k

N

U

I

xt x t













 

0

1

0

1

() , (,)

(4.32)

where x t FI k n ( ,) ( ) represents the interference waveform created by the nth mobile station in

interfering cell k, defined over one time slot only. x t FI k n ( ,) ( )is given by Equation (4.6), which as a

function of , n, and k is written as:

x t ! E

T j tj FI k n

S

( ,) ( ,) , ( ,) FI k n FI k n ( , ) exp ( , ) ,    


2

 0 (4.33)

37
and   (,) k n represents the symbol stream from the nth mobile station in cell k.  FI k n t ( ,) (, )  is

defined over one time slot as a continuously modulated phase based upon the transmitted symbol

stream   (,) k n , and  FI k n ( ,) is a random phase term that is constant over the entire time slot.

 FI k n t ( ,) (, )  may be computed using any of the techniques described in Section 4. The phase

corresponding to an entire time slot is:

  FI k n

m

N

u t t mT t T bits

( ,) (, ) ( ) ,    

 1

1

0 (4.34)

where T is the modulated symbol duration, N bits is the number of bits (symbol durations) per time

slot, and Tu is the duration of a time slot. ( )t is given by Equation (4.34), and represents the phase

component due to a single positive ( ) 1  
1 modulated GMSK symbol. Because only two symbol

states exist for GMSK, ( )t and ( )t (corresponding to 1  
1 and 1  1, respectively), the

phase term for the entire time slot may be expressed as time-delayed versions of

  ( ) ( , )| ( ,) t t  FI k n i0 , scaled by either +1 or -1. Note that each symbol spans more thanT seconds,

therefore symbols will overlap.

In Equation (4.32), x t FI k(,) ( ) represents the sum of all interfering mobile stations from the k

th

interfering cell, as seen at the primary cell’s base station, and x t FI ( ) is the aggregate interference

waveform from all NI interfering cells. Note that x t FI k(,) ( ) and x t FI ( ) are defined over one PCS

1900 frame (eight time slots), corresponding to 4.6 ms. A summary of all values in Equations

(4.32)-(4.34) is given in Table 4.1.

Table 4.1. Summary of Reverse Link Parameter Values

Parameter/Function Description Value

Nbits Number of bits (symbol durations) per time slot 156.25

NU Number of time slots per RF carrier 8

NI Number of interfering cells Integer variable

T Modulated symbol duration 6/1.625 x 106 s

Tu Time slot duration NbitsT s

  (, ) k n Symbol stream from the nth mobile station in cell k Random or user-defined

0, ( , ) FI k n Random phase term that is constant over each

interfering mobile station’s time slot

Random or user-defined; suggest uniform

  0 2,  probability density function (pdf)

2E

T

s Maximum transmission level from each mobile

station interferer 2 V; see Section 4.4

c Waveform propagation speed in medium Typically 3 x 108 m/s

a r kn kn kn ,, , (, )  Propagation loss/Doppler shift function Many models possible; see Section 3.3

TB Time-bandwidth product 0.3

38
4.6.2 Downlink Interference Expressions

GSM downlink interference is caused by base stations outside the primary cell transmitting

unwanted energy to mobile stations within the primary cell. Because the phase modulation for the

downlink interferers is identical to the uplink phase modulation, they will have a form similar to

Equations (4.33) and (4.34) of the previous section, with a slightly different interpretation.

Equation (3.2) gives the generic expression for downlink intrasystem interference, repeated for

convenience:

xt xt 
  r

c

a r FI n FI k n

k n

kn kn kn

k

N

( ,) ( ,)

,

,, , () , 



 



 



	 

0

I 



1

. (4.35)

In Equation (4.35), x t FI k n ( ,) ( )represents the interference waveform to the nth mobile station from the

base station in interfering cell k, which is defined over one frame consisting of eight time slots.

x t FI k n ( ,) ( )is given by Equation (4.6), which, just as for the uplink interference waveform, becomes a

function of , n, and k as follows:

x t 
  ! 
  E

T j tj FI k n

s

( ,) ( ,) , ( ,) FI k n FI k n , exp , ,    


2

 0 (4.36)

where   ( ) k represents the symbol stream from the k th interfering cell’s base station to the nth

mobile station in the primary cell. Note that the symbol stream is not a function of n, since all mobile

stations in the primary cell will encounter the same common symbol stream from the k th interfering

cell’s base station.  FI k t ( ) (, )  is now defined over one time frame as a continuously modulated

phase term based upon the transmitted symbol stream   ( ) k , and0, ( ) FI k is a random phase term

that is constant over the entire frame.

 FI k t ( ) (, )  may be computed using any of the techniques described in Section 4. The phase

corresponding to an entire frame is:

  FI k

m

N N

F t t mT t T bits U( ) (, ) ( )    

'

 1

1

0 (4.37)

where T is the modulated symbol duration, N bits is the number of bits (symbol durations) per time

slot, and TF is the duration of a frame. ( )t is given by Equation (4.34), and represents the phase

component due to a single positive 
  1  
1 modulated GMSK symbol. Because only two symbol

states exist for GMSK, ( )t and ( )t (corresponding to 1  
1 and 1  1, respectively), the

phase expression for the entire frame may be expressed as time-delayed versions of

  ( ) ( , )| ( ) t t  FI k i i0 , scaled by either +1 or -1. As in the reverse link, the GMSK phase terms will

39
overlap. From Equation (4.35), x t FI n ( ,) ( ) represents the sum of all N I interfering base station

waveforms as seen by the nth mobile station in the primary cell. A summary of all values in

Equations (4.35)-(4.37) is given in Table 4.2. Many are identical to those described in Section 4.6.1.

Table 4.2. Summary of Forward Link Parameter Values

Parameter/Function Description Value

Nbits Number of bits (symbol durations) per time slot 156.25

NU Number of time slots per RF carrier 8

NI Number of interfering cells Integer variable

T Modulated symbol duration 6/1.625 x 106 s

TF Frame duration NU Nbits T s

  ( ) k Symbol stream from the base station in the k

th

interfering cell Random or user-defined

0, ( ) FI k Random phase term that is constant over each

interfering base station’s frame

Random or user-defined; suggest

uniform  0 2,  pdf

2E

T

s Maximum transmission level from each base

station interferer 80 V; see Section 4.4

c Interference waveform propagation speed Typically 3 x 108 m/s

a r kn kn kn ,, , (, )  Propagation loss /Doppler shift function Many models possible; see Section 3.4

TB Time-bandwidth product 0.3

4.7 Computer Simulation of the PCS 1900 Noise and Interference Environment

As part of the model validation and verification process, a computer simulation of the PCS 1900 N/I

analytical model was developed. Computer simulation provides valuable insight into the nuances

of model implementation in a real-time hardware channel simulator. As an added benefit, the

computer simulation may be used as a tool to study N/I scenarios based on parametric inputs. For

example, a study of cell size vs. interference level may be conducted by iteratively running the

simulation, while increasing the cell size parameter. Power-level impacts may be profiled by

explicitly setting the transmitted power level of each individual interferer.

The simulation code is designed to be flexible and readily adaptable to many types of TDMA-based

PCS systems. Because PCS 1900 uplink and downlink interference waveforms are very similar,

running simulations for each requires only minor changes in the TDMA-based code. The PCS 1900

uplink N/I simulation is described below. Section 4.7.1 describes the methodology used for

computer simulation of interference to the PCS 1900 uplink, and Section 4.7.2 presents

corresponding sample simulation outputs.

40
4.7.1 Uplink Simulation Methodology for Noise and Interference Generation

Simulation of the PCS 1900 uplink interference provided the aggregate interference waveform caused

by nearby interfering cells, as seen at the primary cell’s base station receiver. A few assumptions were

made to simplify the computations, and to randomize certain parameters as follows:

1. All interfering cells within a given iteration of the program are equidistant from the

primary cell’s base station.

2. Each simulation iteration calculates the aggregate waveform due to a single radius

within each layer of interfering cells. For example, the interference waveform from cells

adjacent to the primary cell will be computed in one iteration, since all cells are

equidistant. Since two different distances are needed for the second layer of interfering

cells, two iterations are required. The total waveform is computed by adding several

simulation iterations together. Obtaining the interference waveform due to interferers in

both adjacent- and second-layer cells therefore requires three simulation iterations.

3. All interfering cells transmit on the same RF carrier frequency. Cells not contributing to

the overall interference waveform (e.g., resulting from a cell reuse pattern) may be

“zeroed out” during the computation.

4. PCS 1900 frames from interfering cells are aligned (this may be changed as a simulation

parameter).

5. Symbols are not aligned from interferer to interferer. A random U T 
 ! 0, delay is

incorporated in each symbol stream.

The methodology for computing the aggregate uplink interference waveform is not complicated,

although the details are rather involved. The major steps, in order of execution, are as follows:

1. Load all parameters for the simulation. These parameters are given in Table 4.3 with

corresponding descriptions and values.

Table 4.3. Uplink Interference Waveform Simulation Parameters

Parameter/Function Description Value

fc

Channel carrier frequency; included as a parameter for

frequency-dependent channel models Ranges from 1850 to 1910 MHz

Pav Average (maximum) transmit power of each mobile station 2 W

symbol Number of possible symbol values; for GMSK, two

symbol values are possible 2

41
Parameter/Function Description Value

T Modulated symbol duration 6/1.625 x 106 s

Ns

Total number of modulated symbol intervals, including

guard times, per time slot 156.25

Ng Total number of guard symbol intervals per time slot

(assumes NORMAL GSM burst type) 8.25

Nu Number of users per frame in primary cell uplink waveform 8

N I u

Number of users per frame in each nearby cell’s

interference waveform 8

c Waveform propagation speed in medium 3 x 108 m/s

R Cell radius 500 m

dxk

Linear X distance of each interfering cell’s base station

with respect to the primary cell’s base station

1000 m (depends on R and the

geographical location of the

interfering cell)

d yk

Linear Y distance of each interfering cell’s base station

with respect to the primary cell’s base station

1000 m (depends on R and the

geographical location of the

interfering cell)

Nic Number of interfering cells

Varies according to the iteration.

Equals 6 for adjacent cells, but

may be set to any number

a r kn kn kn ,, , (, )  Propagation loss/Doppler shift function; may also be set

explicitly to demonstrate a particular power distribution

Default is the simple 1/r4

propagation loss law

2. Load the precomputed phase expression for a single PCS 1900-GMSK symbol. A

separate program was written to precompute the the phase expression from a single PCS

1900-GMSK symbol, as described in detail in Section 4.3.2. This expression is defined

symmetrically over 4T sec., and contains 1000 samples. Since the phase expression is

highly oversampled, it is first decimated by the parameter decfactor. This reduces the

computational load considerably. The decimation factor also defines the simulation

bandwidth for the N/I waveform.

3. Filter the phase expression (optional). The phase expression may be filtered to meet

specified transmission emissions limits. Several different types of filtering are provided

in the program.

Steps 4-6 are used to form a reference desired uplink waveform for the primary cell. This waveform

is not part of the aggregate interference waveform.

4. Calculate positions of users in the primary cell. Mobile stations are randomly placed in

the primary cell. The base station is assumed to be at the origin. Default distribution of

mobile stations is random uniform. In polar coordinates, this means that the angular

42
distribution is U  0 2,  , and the distance from the base station has pdf 2

2

r

R

. The

coordinates for each mobile station are stored in polar form.

5. Calculate propagation loss/Doppler shift function for each mobile station. Any

propagation loss/Doppler shift function may be implemented in this step. The default is

a simple 1

4 r propagation loss law.

6. Calculate the burst from each user. This is accomplished by looping over each time slot

of the PCS 1900 frame. In the process, a random symbol sequence is created for each

time slot. The sequences are modulated in complex baseband form, then shifted and

superimposed to form a single PCS 1900 frame. In the process, the propagation loss

function is incorporated, and guard times included. Modulated symbol streams from

each mobile station include a randomU  0 2,  phase term, and randomU T  0, delay.

In steps 7-10, the aggregate interference waveform is computed. The interference component from

each interfering cell is computed iteratively, then all the components are combined.

7. Calculate positions of mobile stations in each interfering cell. Interfering mobile stations

are randomly placed in the k th interfering cell. Default distribution of mobile stations is

random uniform, with angular distributionU  0 2,  , and base station distance r with pdf

2

2

r

R . Translation of coordinates is accomplished in accordance with the expressions derived

in Section 3.3. The coordinates for each mobile station are stored in polar form.

8. Calculate propagation loss/Doppler shift function for each mobile station in the kth

interfering cell. Again, any propagation loss/Doppler shift function may be

implemented in this step. The default is a simple 1

4 r propagation loss law.

9. Calculate the burst from each user in the k th interfering cell. This is accomplished by

looping over each time slot of the PCS 1900 frame. Similarly, a random symbol

sequence is created for each time slot. The sequences are modulated in complex

baseband form, then shifted and superimposed to form a single PCS 1900 frame.

Equations used were derived in Section 4.6.1. In the process, the propagation loss

function is incorporated, and guard times are included. Modulated symbol streams from

each mobile station include a randomU  0 2,  phase term and randomU T  0, delay.

10. Compute the aggregate waveform. Steps 7 - 9 are executed for each interfering cell.

After each iteration k, the k th interference component is added to the aggregate

interference waveform, which is defined over one PCS 1900 frame. This is expressed

analytically by the second part of Equation (4.32).

43
11. Compute the interference waveform statistics. These include spectral plots, voltage

envelope histograms, power envelope histograms, and phase distributions.

4.7.2 Example Results

Many different N/I scenarios may be created by setting the simulation parameters accordingly, and

consequently a large number of different simulation studies may be conducted. As an example,

simulations were run for the default values listed in Table 4.3. Three simple cases were chosen to

demonstrate the capabilities of the simulation tool:

Case 1: Six interfering cells, all adjacent to the primary cell.

Case 2: Twelve interfering cells, all equidistant from the primary cell.

Case 3: Fifty interfering cells, all equidistant from the primary cell.

For cases 2 and 3, significant overlap in coverage area occured, while case 1 reflects a more realistic

scenario. Three plots were created for each case, showing statistics of the aggregate interference

waveform in the following order: 1) the voltage envelope histogram, 2) the power envelope

histogram, and 3) the phase distribution histogram.

Results for Case 1. Figures 4.7, 4.8, and 4.9 show the voltage envelope, power envelope, and phase

histograms, respectively, of the interference waveform caused by six adjacent interfering cells. The

amplitude envelope distribution appears Rayleigh-distributed, although it is quite distorted in form.

The large value at zero is caused by the guard times, and the simulation assumption that frames in

adjacent cells are aligned (when frames are aligned, so are the guard times—resulting in a large

number of zero values). Randomization of frame alignment in the simulation simply involves

circularly rotating each interfering cell waveform component before aggregation.

Similarly, the power envelope distribution appears as a distorted exponential in Figure 4.8. The

phase distribution of Figure 4.9 is roughly uniformly distributed on the interval   , . Again, the

large count value at zero is caused by the guard times.

Results for Case 2. In Figures 4.10, 4.11, and 4.12, the number of simulated interfering cells was

increased to twelve, with all other parameters the same as for the other cases. Notice that the

amplitude envelope profile given in Figure 4.10 appears more Rayleigh-distributed, the power

envelope in Figure 4.11 appears more exponential, and the phase distribution in Figure 4.12 appears

more uniform than those for case 1.

44
Results for Case 3. The results for the fifty interfering cells of case 3 demonstrate further

smoothing of the distributions, as shown by Figures 4.13, 4.14, and 4.15.

Figure 4.16 shows a sample unfiltered aggregate interference waveform spectrum. Longer samples

of the interference waveform were derived by concatenating numerous instances of the single

frame generated by the simulation.

In summary, this simple case has demonstrated the trend towards a Gaussian interference

waveform as the number of approximately equidistant (equipower) interferers increases. If the

equipower assumption holds, the interference environment for the PCS 1900 uplink may be

modeled as Gaussian noise. However, if the equipower assumption—indirectly specified by the

transmit power of each interfering mobile station—does not hold, what are the consequences?

By setting the power levels explicitly for each interfering mobile station, it is readily demonstrated

that the interference waveform is not purely Gaussian. Figure 4.17 shows the resulting distorted

amplitude envelope when a single interfering mobile station has its transmitted waveform multiplied

by a factor of ten. The distribution and power level of interferers may be explicitly set to any value.

45
46

Figure 4.7. Voltage envelope histogram for case 1.

Figure 4.8. Power envelope histogram for case 1.
47

Figure 4.9. Phase envelope histogram for case 1.

Figure 4.10. Voltage envelope histogram for case 2.
48

Figure 4.11. Power envelope histogram for case 2.

Figure 4.12. Phase envelope histogram for case 2.
49

Figure 4.13. Voltage envelope histogram for case 3.

Figure 4.14. Power envelope histogram for case 3.
50

Figure 4.15. Phase envelope histogram for case 3.

Figure 4.16. Unfiltered aggregate interference waveform spectrum for case 3.
51

Figure 4.17. Voltage envelope histogram for case 1, with a single dominant interferer.

5. IS-95-BASED CDMA-PCS INTERFERENCE WAVEFORM

5.1 CDMA-PCS Physical-layer Overview

Another PCS system targeted for use in the licensed 2-GHz PCS frequency band is the proposed

IS-95-based CDMA, also called CDMA-PCS. Like PCS 1900, CDMA-PCS promises a wide range of

digital services for public large cell applications, including both voice and data. Both systems offer

similar digital user services, but have different implementations on the physical layer, which result in

different performance in the N/I environment. All interference model derivations for CDMA-PCS

described in this report are based on the letter ballot version of the CDMA-PCS standard [21].4

Physical-layer specifications for CDMA-PCS are based on CDMA, which allows many

communication links to simultaneously occupy an RF channel in both time and frequency

dimensions. Unlike TDMA systems, the number of concurrent links occupying a given CDMA RF

channel is not strictly limited. However, practical limits do exist for CDMA capacity.

Self-interference created by several CDMA-PCS transmitters occupying the same RF resource

causes reduced capacity. Links are distinguished by pseudorandom codes that possess strong

autocorrelative properties, and weak cross-correlative characteristics. Ideally, a receiver in a CDMA

system will have perfect correlation with its assigned link, and all other links sharing the RF channel

will appear as noise. This depends on the design of the CDMA codes, and the system deployment.

Unlike its PCS 1900 counterpart, CDMA-PCS employs distinctly different designs on the forward

and reverse links. Commonality exists, however, in the chipping sequence rate of 1.23 Mchips/sec.

This chipping sequence, after modulation, occupies a 1.23-MHz bandwidth. Logical channels are

distinguished by two parameters: 1) the RF carrier frequency and 2) the logical channel-specific

spreading code. Data rates up to 14.4 kbps are provided per logical channel.

5.1.1 Reverse Link Design

On the reverse link (uplink), two types of logical channels are specified: 1) access channels and 2) traffic

channels. Access channels are used by the mobile station to establish traffic channel links with the base

station, and to respond to base station signaling. Traffic channels communicate voice, data, and

signaling information to the base station from the mobile station. For N/I modeling purposes, the traffic

and access channels are assumed identical, except for the private spreading codes.

53

4 Document number T1P1/94-088 (J-STD-008), released 12/12/94. The ballot version

specifies technical requirements for CDMA-PCS compatibility.
Figure 5.1 shows the reference configuration for the CDMA-PCS reverse link, which assumes a

9.6-kbps logical traffic channel. Other logical channel rates and the access channels will have a

slightly different configuration. This includes error-coding, block-interleaving, and symbol

repetition, which do not directly contribute to the N/I waveform in the model.

Shaded blocks in Figure 5.1 represent functions which may be explicitly represented in the N/I

model. The 64-ary Walsh code modulator takes groups of 6 symbols from the interleaver, and maps

each group into one of 64 Walsh codes. Each Walsh code is 64 “chips” long. These are passed to a

burst randomizer at 307 kchips/sec. The burst randomizer masks out redundant data generated by

the symbol repeater. For N/I modeling purposes, this function is ignored. The Walsh codes are then

xor’ed with a private pseudorandom code which is 242 -1 chips long. In the process, each Walsh

chip is modulo-2 added to four long code chips, resulting in a chip stream at 1.23 Mchips/sec. The

long code distinguishes individual users within the RF channel resource. Although ultimately the

long code is the same for every user, a sufficient “offset” in the code, resulting in near-othogonal

properties, is emulated by proper application of a user-specific privacy mask. The long code has

sufficient length to support many concurrent offsets.

54

Information

bits in Frame Quality

Indicator

Add 1 byte

encoder tail

to frame

Convolutional

Encoder

Symbol

Repeater

Block

Interleaver

Walsh Code

64-ary

Modulator

Burst

Randomizer

Long Code

XOR

I and Q

XOR

OQPSK

Modulator

To air

interface

Figure 5.1. CDMA-PCS reverse link block diagram.
After long code-spreading, the symbol stream is split into I and Q components, which are modulo-2

added synchronously to corresponding I and Q spreading codes of length 215 each. The I and Q

codes are the same for every mobile station sharing the same RF resource for a given cell. Nearby

cells also make use of the same I and Q codes, but use a time offset sufficient for decorrelation with

other nearby cells. Offset quadrature phase-shift keying (OQPSK) is then used to modulate the I

and Q symbol streams, which are then transmitted over the air interface.

Functions that do not affect the N/I waveform directly are omitted in the model. However, more

detail is needed in modeling CDMA-PCS because the correlation properties of the N/I waveform

directly impact the performance of the correlator-based receiver. For this reason, the model allows

direct implementation of the spreading codes used by the interferers in the aggregate waveform. As

an alternative, random chip generators may be substituted for the actual chip sequence generators.

This reduces the computational complexity of the N/I waveform considerably. Figure 5.2 shows the

resulting abstraction that applies to the reverse link waveform.

5.1.2 Forward Link Design

For the forward link (downlink), four types of logical channels are defined: 1) the pilot channel, 2) the

sync channel, 3) paging channels, and 4) traffic channels. The pilot channel is simply a spread spectrum

waveform modulated by Walsh function 0 that serves as a synchronization beacon for mobile stations

within range of the RF carrier. The sync channel provides timing information for each mobile station,

once pilot channel synchronization is established. Paging channels are used to communicate system

overhead information and signaling messages from the base station to the mobile station.

Forward link design for CDMA-PCS differs significantly from the reverse link in modulation

scheme, synchronization, and spreading code methodology. Because all forward link logical

channels are combined within the base station, all are synchronized. For this reason, a mobile

station may extract all timing information from the base station pilot and sync channels, and

demodulation is therefore accomplished coherently. Forward link Walsh functions are used to

distinguish logical channels, in contrast to orthogonal modulation implementation on the reverse

55

Information

bits in All spreading

codes XOR Transmitter OQPSK

Modulator

To air

interface

Figure 5.2. Abstracted reverse link block diagram.
link. Figure 5.3 shows the reference configuration for the CDMA-PCS forward link, which again

assumes a 9.6-kbps logical channel.

Shaded blocks in Figure 5.3 represent functions that may be explicitly represented in the N/I model.

These blocks directly affect the correlative properties of the mobile station receiver. The symbol

stream output from the block interleaver is modeled as a random bit stream, which is then modulo-2

added to a logical-channel-specific long code decimated by a factor of 64. The symbol stream is

then multiplexed with power control information and a version of the long code decimated by a

factor of 1536. The multiplexed symbol stream is then modulo-2 added to the

logical-channel-specific Walsh function, then I and Q spread at 1.23 Mchips/sec before

modulation. The forward channel uses QPSK without an offset.

Although the amount of processing in the forward (and reverse) CDMA-PCS links seems prohibitive

for an explicit N/I waveform simulation with many interferers, many of the blocks may be simplified

by aggregating them together. For example, if sufficient randomness is built into the symbol stream

output from the block interleaver, the remaining shaded blocks—with the exception of the

56

Information

bits in Frame Quality

Indicator

Add 1 byte

encoder tail

to frame

Convolutional

Encoder

Symbol

Repeater

Block

Interleaver

Decimated

Long Code

XOR

Walsh

Function

XOR

QPSK

Modulator

To air

interface

Symbol

Puncturer

MUX code,

power control,

and data

I and Q

XOR

Figure 5.3. CDMA-PCS forward channel block diagram.
modulator—may be replaced by an equivalent code spreading function. This code-spreading function

is based on the actual codes used in the CDMA-PCS system. Figure 5.4 shows this abstraction.

To further simplify the N/I simulation, random number generators can replace the actual chipping

sequence generators. This substitution will not capture the correlative properties of the actual

pseudorandom codes, but works well if the codes are near-orthogonal.

5.2 Offset QPSK Waveform Expression (Reverse Link)

Mobile station modems implement OQPSK on the reverse link. The mobile station output symbol

stream, after the xor function with the mobile station-specific long code, is separated into in phase and

quadrature symbol streams for modulation. The I and Q separation provides diversity in the

transmitted symbol stream. Unlike traditional uses of quadrature modulation that alternate successive

bits before transmission, CDMA-PCS I and Q symbol streams each carry the same information.

In CDMA-PCS OQPSK, the modulator power divides the long-coded symbol stream into two

identical components, then spreads each component with corresponding I and Q pseudorandom

sequences. The Q symbol stream is delayed by one-half a chip duration to form the offset. Next, I

and Q symbol streams are baseband filtered, then quadrature modulated. The complete reverse link

process is shown in Figure 5.5.

57

Information

bits in All spreading

codes XOR Transmitter QPSK

Modulator

To air

interface

Figure 5.4. Abstracted forward link block diagram.



Low-Pass

Filter

Low-Pass

Filter delay

I symbol stream

Q symbol stream

I code

Q code

symbol

stream

I(t)

Q(t)

x(t)

cos(2 ft) 

90 phase

shift

°

Figure 5.5. Offset quadrature phase-shift keying for the CDMA-PCS reverse link.
In terms of the generalized phase-modulated waveform given by Equation (A.3), the complex

baseband representation of a single transmitted OQPSK waveform is given by Equation (5.1):

x t ! E

T j t s ( , ) exp ( , ) .    2

 (5.1)

In Equation (5.1),(, ) t  is the continuous information-carrying phase expression. For OQPSK, the

phase expression (, ) t  will transition between four states as shown in Figure 5.6. Note that

transitions between state pairs 1,3 and 0,2 never occur. This is a consequence of offsetting the Q

symbol stream. As a result, instantaneous phase changes are limited to  / 2 radians.

The simple four-state model shown in Figure 5.6 represents an idealized OQPSK modulator, with

instantaneous transitions between phase states every T/2 sec., where T is the chip duration. More

realistically, baseband-shaping of the I and Q components of the phase determines the spectral limits

of the output waveform. In effect, baseband-shaping creates a continuum of intermediate “states”

between the four states shown in Figure 5.6, eliminating the instantaneous transitions entirely.

For N/I modeling purposes, many methods may be used to emulate the phase term (, ) t  ; two are

discussed in this report. One approach is to use a Markov chain model to emulate the phase state

changes, and filter the entire phase expression after modulation. Another approach is the explicit

modeling of the I and Q symbol streams, which are then baseband filtered and mapped into the

phase expression. This essentially involves the same processing as the real reverse link, shown in

58

)/4

-3 /4  /4

/4

State 1 State 0

State 2 State 3

Figure 5.6. Offset quadrature phase-shift keying state transition diagram.
Figure 5.5. The approach used depends on the amount of detail desired in the N/I simulation, and

the computational simplicity desired. Both approaches are described in the following two sections.

5.2.1 Markov Chain Model

Markov chain models offer many computational advantages over “brute force” methods of

generating an unfiltered random OQPSK waveform from a single modulator. This is because each

phase transition is represented by a single data point. This corresponds to two phase transitions per

chip duration, or approximately 2.46 Msamples/sec/modulator. While this appears large, for

filtered versions using I and Q symbol streams, each stream must be sampled at the simulation

bandwidth, which is typically 4-16 times the RF bandwidth. This results in a sampling rate of

approximately 5-20 Msamples/sec for each of the two output symbol streams per modulator. Each

symbol stream must then be filtered separately before transmission.

Unfortunately, there are also many disadvantages to using the Markov chain model. For each

additional modulator, the number of samples necessary increases linearly until the simulation

bandwidth is reached. This is due to the noncoherent nature of each mobile station transmission, and

because each transmitted symbol stream will have a random time offset with respect to other

transmitting mobile stations. In addition, filtering the Markov chain output according to CDMA-PCS

specification is difficult, and requires a sampling rate at the simulation bandwidth. Also, because the

Markov chain model is stochastic, the correlative properties of the spreading codes are not taken into

account. For these reasons, the Markov chain model is not used in the ITS N/I model.

Despite the disadvantages associated with the Markov chain model, the methodology is presented here

for those who wish to use it. Although not practical for modeling large numbers of interferers, the

Markov chain model may prove useful for single-interferer analyses, and for modem-emulation studies.

The Markov chain model is based on the state diagram shown in Figure 5.6. Defining  n m, as the

probability of transitioning from state n to state m, the transition matrix for the state diagram is

defined as:









00 01 02 03

10 11 12 13

20 21 22 23

,,,,

,,,,

,,,,



 



30 31 32 33

00 01 03

1 0

0

,,,,

,, ,

,



































 



 

11 12

21 22 23

30 32 33

0

0

0

, ,

,,,

, ,,

































, (5.2)

59
where the zero probability transitions are caused by the half-chip offset in the Q symbol stream. The

phase states are related to the I and Q symbol values as summarized in Table 5.1.

Table 5.1 OQPSK Phase State Mapping

State Phase Value I Symbol Value Q Symbol Value

0 / 4 0 0

1 3 4 / 1 0

2 3 4 / 1 1

3 / 4 0 1

Transitions from state i to state i+1 occur at times nT

2 , where n is the set of non-negative integers. If

all symbols are assumed equiprobable, the transition matrix becomes:



































1

3

1

3

1

3

1

3

1

3

1

3

1

3

1

3

1

3

1

3

1

3

1

3

0

0

0

0

, (5.3)

Equation (5.3) can be confirmed by solving the steady-state probability vector equation p p  ,

where the sum of elements in the vector p  ! pppp 0123 equals unity, and p j is the probability

that the Markov chain is in state j. The resultant p j are all equal to 0.25, indicating that all states are

equiprobable.

From the perspective of the primary cell’s base station, the interference waveform component from

each mobile station is received with a random phase that is constant over the transmission, and a

random time offset that is proportional to the distance of the mobile station from the base station.

Because the reverse link phase is inherently noncoherent at the base station receiver, the phase

component from each mobile station will be determined by separate, independent, and identical

Markov chains. For computational purposes, the states of the individual Markov chains may be

represented compactly and updated via the following steps:

1. Define a Markov “flag” matrix M of dimension {Nu, S}, where Nu is the number of

interfering mobile stations, and S is the integer number of phase states. Matrix M stores

the phase transitions of each interfering mobile station. Rows of M identify individual

mobile stations, and columns indicate the current phase state. A “1” in a column

indicates the current phase state. For OQPSK, S=4.

60
2. Choose initial states for M. This can be accomplished by randomly choosing a state for

each mobile station. This means one column for each row of M is randomly set to “1.”

3. Choose a random phase constant for each row of M. The n

th mobile station waveform

will arrive at the base station with a constant random phase 0n . A uniform  0 2, 

distribution is appropriate.

4. Form a phase state equation that stores the phase of each mobile station. Define

s  ! sss s 012  s , where si is the value of the i

th state, and

0 00 01 02 0     !  N u ; and where i is the current phase value, including the

random offset, of the i

th mobile station. Also, let    0 00 01 02 0  !  N u

represent the random phase constant matrix, where0i is the random phase offset of the

i

th mobile station. A phase state equation can then be formulated as:   
 sM T 0 .

5. Update the M matrix. States of the M matrix will be updated for every

mobile-station-initiated change of phase. For OQPSK, the state of each row will be

updated every T/2 sec. Because time delays are associated with propagation for each

mobile station, a random time offset uniformly covering a single chip duration is

included. Thus, each row of the M matrix is updated twice every T sec., resulting in 2Nu

updates per chip duration for the entire interference waveform. M can be updated by:

a. Defining a {1, Nu} time vector T with elements distributedU T  0 2 , / .

b. Ordering the elements of T from smallest to largest.

c. Updating consecutive rows of M at times specified by the elements of T. For

example, row M0 is updated at time T0, M1 is updated at time T1, and so forth. The

state of M can be updated for OQPSK by only observing columns (j-1, j, j+1)

modulo S, where j is the current state, and choosing state (j-1, j, j+1) modulo S

with probability 1/3.

d. Add T/2 to all elements of T when all rows of M have been updated, then

proceed as in step 5c.

6. Update the phase state equation in step 4.

Obviously, the Markov chain method becomes cumbersome for large numbers of interfering

mobile stations.

61
5.2.2 I and Q Explicit Model

An alternative method of OQPSK waveform generation uses an explicit model of the modulator, as

shown in Figure 5.5. While this method is more computationally intense than the Markov chain

model for small numbers of interferers, the many advantages make this the preferred method for the

ITS N/I model. Explicit modeling allows inclusion of the actual spreading codes, in addition to

direct application of baseband filtering used in the real system design. If desired, abstractions are

easily incorporated that simplify the model further (e.g., use of a random number generator in place

of a spreading code generator). In addition, for large numbers of interferers, the computational

complexity is actually less than a comparable Markov chain model.

Computational complexity for the I and Q explicit model is based on the simulation sampling rate,

which is typically 4-16 times the baseband bandwidth of the RF waveform. For an explicit model,

the simulation sampling rate is independent of the number of interferers, because all waveforms are

sampled at identical times, i.e., synchronously. For CDMA-PCS, the simulation sampling rate

typically ranges from 5-20 Msamples/sec. For the N/I model, the lower sampling rate is used.

Modeling the N/I waveform explicitly is straightforward. The 1.23-Mchip/sec input symbol stream,

after modulo-2 addition to the long code, is power divided into I and Q components, which are then

modulo-2 added to corresponding I and Q codes. The Q symbol stream is delayed by one-half a chip

duration, and then both I and Q streams are placed through the lowpass filter. The streams are next

modulated in quadrature. This process, in complex baseband form, is shown in Figure 5.7.

62



h(t)

delay h(t)

I (t) UF I (t) F

Q (t) UF Q (t) F

Q symbol stream

I code

Q code

symbol

stream

I(t)

Q(t)

x(t)

j

XOR

XOR

1

Figure 5.7. CDMA-PCS OQPSK modulator, in complex baseband form.
The modulator output in Figure 5.7 is the sum of in phase and quadrature components:

xt I t ft Q t ft F c F c ( ) ( )cos( ) ( )sin( ),  
 2 2   (5.4)

which is in canonical form. The complex baseband representation is simply:

x t I t jQ t F F ( ) ( ) ( ).  (5.5)

In terms of the unfiltered I and Q symbol streams, the complex baseband output is given by the

convolution:

x t h t I t jQ t T

UF UF

c () () () .  



 



	 









 2 (5.6)

The complex baseband output x t( ) is simply the sum of I and Q symbol streams, with appropriate

linear filtering. Filter limits are given in [21]. Details of the technique used for simulation of the

reverse link modulation is given in Section 5.7, including sampling method, code implementation,

and filter design.

Abstraction may be included during many steps of the process. For example, the input symbol

stream may be replaced by a random number generator, at the cost of losing the correlative

properties of the Walsh and long codes. I and Q spreading codes may be replaced similarly.

5.3 QPSK Waveform Expression (Forward Link)

Base station modems implement baseband-filtered QPSK on the forward link. The modulation

scheme is identical to that for the reverse link, except the Q symbol stream is not offset.

As in the reverse link, the CDMA-PCS QPSK modulator accepts the long-coded symbol stream,

divides it into two identical components, then modulo-2 adds each component with corresponding I

and Q pseudorandom sequences. For the forward link, the Q symbol stream is not delayed. I and Q

symbol streams are next baseband filtered, then quadrature modulated. The complete forward link

process is shown in Figure 5.8. The complex baseband representation of a single transmitted QPSK

waveform is given by Equation (5.7):

x t ! E

T j t s ( , ) exp ( , ) ,    2 (5.7)

63
where   (, ) t is the continuous information-carrying phase expression. This expression is identical

to the phase expression given for the reverse link in Equation 5.1. For QPSK, the phase expression

  (, ) t will transition between four states as shown in Figure 5.9.

64



Low-Pass

Filter

Low-Pass

Filter

I symbol stream

Q symbol stream

I code

Q code

symbol

stream

I(t)

Q(t)

x(t)

cos(2 ft) 

90 phase

shift

°

Figure 5.8. QPSK for the CDMA-PCS forward link.

)/4

-3 /4  /4

/4

State 1 State 0

State 2 State 3

Figure 5.9. QPSK phase state transition diagram.
Unlike OQPSK, transitions between state pairs 1,3 and 0,2 do occur. Phase changes happen only

once per chip, since the I and Q symbol streams are aligned. However, instantaneous phase changes

of radians result, which are more difficult to manage in an actual transmitter/receiver pair.

The phase term(, ) t  may be emulated via any of several methods, as was the case for the reverse

link. Both the Markov chain model and the I and Q explicit model described in Section 5.2 apply,

with some minor changes. The methodologies are the same for each case. Rather than repeat each

methodology, only the changes are identified below.

For the Markov chain model, the transition matrix becomes:









00 01 02 03

10 11 12 13

20 21 22 23

,,,,

,,,,

,,,,

 30 31 32 33 ,,,,

.

































(5.8)

Note that the zero transition probabilities disappear, although the phase values in Table 5.1 still

apply. If all symbols are assumed equiprobable, all the transition probabilities are equal to 0.25.

Computation of the phase state equation is the same, except the phase state of each individual

interferer is updated every T sec., instead of every T/2 sec.

For the I and Q explicit model, Equation (5.6) becomes:

x t h t I t jQ t ! UF UF () () () ().  (5.9)

5.4 CDMA-PCS Power Control

An important element of any CDMA system design is power control on the reverse link. An

improperly balanced CDMA system will suffer significant losses in capacity. A well-known case of

CDMA system imbalance is referred to as the “near/far” problem, where a mobile station “near” the

base station receiver transmits well above the levels of other “far” away mobile stations on the same

RF channel. Of course, the power levels received at the base station from each mobile station may

depend on factors other than geographic distribution. More generally, the mobile station transmit

powers will be distributed in envelope in a time-variant manner. The ITS N/I model allows

specification of the transmit power of each mobile station in the overall N/I waveform.

65
The CDMA-PCS system uses both open and closed loop power control. For open loop power

control, the product of the power received at the mobile station and the power transmitted from the

mobile station is a defined constant. The farther away the mobile station is with respect to the base

station, the greater the allowed mobile station transmit power. Specifically, the output power

characteristics of the mobile station (traffic channel) are governed by formulae for initial power

transmission level, and open loop estimation during a call session. A change in the mean input

power level to the mobile station will trigger a proportional change to the output power level in the

opposite direction. Open loop power control is governed solely by the mobile station.

In addition, a closed loop correction controlled by the base station allows mobile station power

adjustments to ±24 dB from the open loop estimate. Closed loop power control bits are transmitted

from the base station periodically, and instruct the mobile station to increase or decrease

transmitted power levels by a nominal increment (typically 1 dB).

Rather than implement the CDMA-PCS power control algorithm directly in the uplink N/I model,

functionality is included to choose a distribution of power levels from interferers. Weighting

functions, which may be functions of time, allow specific setting of interferer levels. In theory,

these functions may emulate the power control algorithm used in CDMA-PCS, but a more practical

approach is to develop stationary scenarios for the distribution. In addition to being much simpler to

implement, a stationary distribution facilitates easier “what if” analyses related to power control.

Example specification of the weighting functions are given in the example in Section 5.7.1.

5.5 Timing and Synchronization

Another important consideration for the N/I model is understanding timing and synchronization for

the CDMA-PCS system. Resulting effects that are implemented in the N/I model are 1) phase

synchronization of logical channels on the forward link, 2) noncoherent reception of logical

channels on the reverse link, and 3) implementation of spreading code offsets.

CDMA-PCS uses a universal time reference based on global positioning system (GPS)

transmissions to achieve system synchronization. Long, I, and Q codes are synchronized at the

transmitter and receiver via the GPS transmissions. Mobile stations acquire synchronization and

system time via the following steps:

1. Mobile stations, upon entering the CDMA-PCS environment, scan for the base station

pilot channel. The pilot channel is a continuous transmission of Walsh code 0.

66
2. Once the pilot channel has been located, the mobile station acquires timing and phase

references for coherent demodulation. Phase synchronization is maintained between all

channels on the forward link, making coherent demodulation possible.

3. After acquisition of the pilot channel, the mobile station uses the base station sync

channel to obtain system configuration and timing information. Included are the

tracking and timing offset information needed for proper decorrelation of the long, I,

and Q spreading codes.

For the ITS N/I model, all logical channels on a single RF channel for any base station on the

forward link are assumed to be synchronized. RF channels from nearby interfering base stations are

not synchronized with respect to each other, and each uses a different I and Q spreading code offset.

On the reverse link, all mobile station transmissions are assumed to have a randomU ,  0 2 phase

offset and a random U T  0, delay, as received at the base station. However, all mobile stations

assigned to a given base station use the same I and Q spreading code offset, not including

propagation delay. Mobile stations in nearby interfering cells use the cell-assigned I and Q

spreading code offset.

5.6 Interference Expressions

Section 3 presented the PCS cellular geometry of intrasystem interference in a generic,

technology-independent form. Section 5, to this point, has described the waveform created by a

single modulator on both the uplink and downlink. CDMA-PCS-specific N/I environments may be

formulated using expressions derived in both sections. Single-modulator waveforms for the uplink

and downlink can be substituted into the geometric expressions presented in Section 3 to

characterize the CDMA-PCS system self-interference.

The methodology is similar to that used for the PCS 1900 self-interference characterization. Two

cases are presented: 1) interference to the uplink and 2) interference to the downlink. For each case,

numerous interferers are simulated by duplicating a single transmitter expression once for each

represented interferer. Each interferer is then spatially distributed via the expressions derived in

Section 3. Delay, multipath, propagation loss, and Doppler shift functions may also be incorporated

in these expressions.

67
5.6.1 Uplink Interference Expressions

On the uplink, CDMA systems inherently experience intracell interference from other mobile

stations within the cell, in addition to mobile stations transmitting on the same frequency in nearby

cells. In general, each mobile station transmits a waveform into the waveform space shared by all

other mobile stations assigned to the same RF channel, independent of cell assignment. Individual

waveforms are discernable by code assignment.

Unlike TDMA-based systems, CDMA has a soft capacity that depends on several environmental

parameters. Specifically, the intracell component of the self-generated CDMA-PCS uplink N/I

waveform for a single RF channel will be a function of the:

1. number of mobile stations concurrently transmitting on the RF channel,

2. positions of the mobile stations with respect to the primary cell’s base station receiver

and each other,

3. channel conditions and propagation environment,

4. transmitted power distribution of the mobile stations and power control algorithm, and

5. design of spreading codes (correlative properties).

An appropriate N/I model for the CDMA-PCS uplink accounts for these factors, with facility for

abstraction where possible. Note that these factors are not independent. For example, the positions

of mobile stations within the primary cell determine their individual transmitted power. Assuming

perfect power control, all mobile station waveforms incident at the base station receiver have equal

power levels. Obviously, mobile stations near the border of the primary cell must transmit at a

higher power level than those near the base station to accomplish this. Thus, the power distribution

of mobile station waveforms measured at the mobile station transmitters is related to the position of

the mobile stations.

Equations (3.1) give the technology-independent expressions for uplink intrasystem interference.

The parameter b is set to zero in Equations (3.1), allowing superposition of multiple CDMA

waveforms in the time domain. Equations (3.1) then become:

xt xt 
  r

c

a r FI k FI k n

k n

kn kn kn

n

N

( , ) ( ,)

,

,, , () , 



 



 



	 

0


 

U

I

xt x t FI FI k

k

N











 

1

0

1

() . (,)

(5.10)

68
Because mobile stations sharing the same RF uplink channel mutually interfere within the primary

cell, the number of interfering cells, NI, must include the primary cell. The k=0 index may

arbitrarily represent the primary cell.

Frame boundaries and time slots are meaningless for the N/I model. Therefore, the interference

component x t FI k n ( ,) ( ) from the n

th interfering mobile station in the kth cell, is defined over an

arbitrary period of time. Although CDMA-PCS uses frame structures, the interference waveform

need not include that level of detail. For convenience, x t FI k n ( ,) ( ) may be defined in terms of an

integer number of chips, or over an entire spreading code cycle. With the understanding that the

time duration is arbitrary, x t FI k n ( ,) ( ) is adapted from Equation (5.1), and expressed in terms of a

single OQPSK modulator as:

x t At j t j FI k n k n k n k n ( , ) , , ,( , ) ( , ) ( )exp ( , ) ,    
 !  0 (5.11)

where   ( ,) k n represents the symbol stream from the n

th mobile station in cell k, ( ,) (, ) k n t  is

defined over one time slot as the continuously modulated phase based upon the transmitted symbol

stream,   ( ,) k n , 0,( , ) k n is a random phase term that is constant for the entire waveform, and

A t k n, ( ) is the power-controlled output amplitude of the waveform.

Section 5.2 presented two methods for computing the output of a single reverse link OQPSK

modulator. Note that Equation (5.11) also represents the waveform at the transmitter output of the

n

th mobile station in the kth cell, with two additional “features.” Specifically, Equation (5.11) adds a

power control function and a random constant phase term. Both are independent of the modulation

scheme. Modulation-dependent components of x t FI k n ( ,) ( ) are computed using Equations (5.1) or

(5.6). Remember that modulation-dependent expressions include all processing before modulation,

including spreading codes. These must be chosen appropriately for every x t FI k n ( ,) ( ). Equation (5.10)

results by taking the modulation-dependent components and incorporating the random constant

phase term and power control function. Mathematically,

x t A txt j FI k n k n k n ( , ) , ,( , ) ( ) ( ) ( , )exp( )  0

for Equation (5.1), and

x t A txt j FI k n k n k n ( , ) , ,( , ) ( ) ( ) ( )exp( )  0

for Equation (5.6).

69
Aggregate interference caused by all mobile stations in the k

th cell, as received by the primary cell’s

base station, is expressed by combining Equations (5.10) and (5.11):

x t At

r

c

j t

r

c

FI k k n

k n

k n

k n

(,) ,

,

,

, ( ) exp ,  



 



	 



 

  
  

	 












 



 j ar kn kn kn kn

n

NU

  0

0

1

,( , ) , , , , . (5.12)

Although a r kn kn kn ,, , 
  , is not explicitly expressed as a time function, it may change value over

time due to fluctuating channel conditions, or changing position of the mobile station. The product

of the two amplitude functions is defined as:

W t A ta r t kn kn kn kn kn , , ,, , () () , , .   
  (5.13)

The new functionW t k n, ( )is directly related to the power control algorithm. By setting W t k n, ( ) constant

for all n and setting k=0, perfect power control is represented within the primary cell. Explicit choice of

the W t k n, ( ) allows different power control profiles to be represented in the N/I model. These can be

either from a random distribution, or manually set according to a specific scenario.

To compute the total uplink interference waveform, Equation (5.12) is summed over all k, as given

in the second part of Equation (5.10).

Rather than provide a table of values for the CDMA-PCS uplink interference waveform

expressions as was done for the GSM uplink interference expressions in Section 4.6.1, an example

is given in Section 5.7.2. This will most likely be more helpful, since the large number of possible

parameter settings and techniques is difficult to cover in a succinct manner.

5.6.2 Downlink Interference Expressions

Forward link logical channels, unlike their reverse link counterparts, are combined before air

transmission within each RF channel. At the base station, each logical channel is separated into in

phase and quadrature components, which are converted to analog waveforms before filtering. Next,

all logical channels are combined in the baseband analog domain, then modulated for air

transmission. Forward link logical channels are thus synchronized, making coherent detection in

the mobile station receiver possible.

Modeling the forward link involves proper synchronization of all logical channels within the RF

channel. Equivalently, the logical channels may be modulated then summed, or summed then

modulated. To illustrate this, Figure 5.10 reflects one possible logical channel-combining

technique at the base station.

70
In Figure 5.10, baseband analog versions of I and Q components, x t iI ( )and x t iQ ( ), from each logical

channel are QPSK-modulated to form N waveforms,s t i( ), at carrier frequency f c . The s t i( )are then

summed and sent to the transmitter. This method employs N QPSK modulators. Another method

for combining the logical channels on the forward link is shown in Figure 5.11. In this example, the

I and Q baseband analog components, x t iI ( ) and x t iQ ( ), are summed separately to form aggregate I

and Q components for the RF channel, x t I ( ) and x t Q ( ). These two components are then QPSK

modulated and sent to the transmitter. Note that only one QPSK modulator is used.

Figure 5.10 yields:

s t x t ft x t ft

st x t

i iI c iQ c

iI

( ) ( )cos( ) ( )sin( ),

() (

 




2 2  

)cos( ) ( )sin( ) 2 2   ft x t ft c

i

iQ c

i

 
 (5.14)

and from Figure 5.11:

xt x t x t x t

st x t f

I iI

i

Q iQ

i

I

( ) ( ), ( ) ( ),

( ) ( )cos(

 



  and

2 c Q c t x t ft ) ( )sin( ). 
 2

(5.15)

Clearly, using either method yields the same result for explicit modeling of the I and Q symbol

streams from each logical channel. The second method is not appropriate if a Markov chain model

is used to model each modulated symbol stream. For N/I modeling purposes, the I, Q, and long

spreading codes may be applied in any order, since the xor function is communative. Note that the

71

x (t)

x (t)

1I

1Q

s (t) 1

s(t)

x (t)

x (t)

NI

NQ

s (t) N

x (t)

x (t)

iI

iQ

s (t) i

x (t)

x (t)

2I

2Q

s (t) 2

*

Figure 5.10. Technique 1 for CDMA-PCS forward channel combining.
real and imaginary parts of the complex baseband output of a single forward channel modulator, as

given by Equation (5.9), correspond to x t iI ( ) and x t iQ ( ), respectively.

Equation (3.2) gives the technology-independent expression for downlink intrasystem interference.

If the multiple QPSK modulator method of Figure 5.10 is used to generate a base station RF

channel, then s t i( ) is the bandpass version of x t(, )  as given by Equations (5.7) and (5.9). Using

complex baseband notation, the interference waveform from the k

th interfering base station, as

measured at the k

th base station’s transmitter is:

x t A t xt FI k n k i k i ! i

N u

( ,) , , ( , ) ( ) ( , )| ,    



1

(5.16)

where i indicates the i

th logical channel component of the waveform from base station k, and x t k i ( , )|  ,

is the ith modulated component in complex baseband form. The index n indicates that the interference

waveform x t FI k n ( ,) (, )  affects the n

th mobile station in the primary cell. Note that this expression is

not dependent on n, since the waveform is assumed at the k

th base station’s transmitter. The n is kept

for notational consistency. The function A t k i, ( ) allows the relative amplitude of each logical channel

to be adjusted. For example, the pilot channel is usually transmitted at a higher level than other logical

channels within the RF channel to facilitate easier detection.

72

x (t)

x (t)

1I

1Q

s(t)

x (t)

x (t)

NI

NQ

x (t)

x (t)

iI

iQ

x (t)

x (t)

2I

2Q

*

*

*

fc

-1

I

Q

x (t) I

x (t) Q

Figure 5.11. Technique 2 for CDMA-PCS forward channel combining.
Substituting Equation (5.16) into Equation (3.2) gives the aggregate downlink interference

expression from NI nearby CDMA-PCS base stations, as seen by the n

th mobile station in the

primary cell. Equation (3.2) is repeated below for convenience:

xt xt 
  r

c

a r FI n FI k n

k n

kn kn kn

k

( ,) ( ,)

,

,, , (, ) , 



 



 



  	

0

N I 1

 . (5.17)

Downlink CDMA-PCS intrasystem interference at the primary cell’s n

th mobile station is caused by

nearby base stations radiating unwanted energy into the RF receive channel, and other downlink

logical channels mutually interfering within the primary cell’s base station channel. Because the

primary cell’s base station creates a waveform with interfering logical channels, the aggregate

interference waveform must include all logical channels transmitted on the primary cell’s base

station downlink channel. Consequently, NI must include the primary cell. The N/I waveform

should ideally be synchronized with the “real” communication forward logical channel, as is the

case for all logical channels in the forward link.

Alternatively, a test scenario may include a mobile station-to-base station link with a fully loaded

forward channel. In this case, NI does not include the primary cell’s base station. However, the

“real” base station must be able to transmit logical channels to mobile stations which may be

required to physically exist in the test bed. If this is the case, registration and call setup procedures

are needed for each mobile station; these are not incorporated into the N/I waveform models.

5.7 Computer Simulation of the CDMA-PCS Noise/Interference Environment

Part of the model validation and verification process for the CDMA-PCS was construction of

computer simulations for the N/I waveforms derived in Section 5. These simulations provide

valuable insight into the nuances of model implementation in a real-time hardware channel

simulator. In addition, the computer simulation may be used as a tool to study N/I scenarios based

on parametric inputs. This is particularly useful for CDMA-based systems, where power control

impacts may be profiled by explicitly setting the transmitted power level of each individual

interferer according to a prederived distribution, or via the CDMA-PCS power control algorithm.

The simulation code is readily adaptable to different CDMA systems. Unlike the PCS 1900 system,

CDMA-PCS forward and reverse links differ in modulation scheme, spreading order, and

synchronization. Fortunately, the parametric inputs to the code allow easy adaptation of the

simulation to either uplink or downlink. To avoid repetition, only the uplink N/I simulation is

detailed in this report.

73
The CDMA-PCS uplink N/I simulation is described below. Section 5.7.1 describes the

methodology used for computer simulation of interference to the CDMA-PCS uplink, and Section

5.7.2 presents corresponding sample simulation results.

5.7.1 Uplink Noise/Interference Simulation

CDMA-PCS uplink interference simulation provides the aggregate interference waveform caused

by mobile stations in the primary cell and nearby interfering cells, as seen at the primary cell’s base

station receiver. Assumptions simplify the computations in a similar manner to those made for the

PCS 1900 uplink interference waveform simulation in Section 4.7.1. In fact, some assumptions are

identical. These are:

1. All interfering cells within a given iteration of the program are equidistant from the

primary cell’s base station. This is a technology-independent assumption.

2. Each simulation iteration calculates the aggregate waveform due to a single radius

within each layer of interfering cells. For example, the interference waveform from cells

adjacent to the primary cell are computed in one iteration, since all cells are equidistant.

Since two different distances are needed for the second layer of interfering cells, two

iterations must be run. The total waveform is computed by adding several simulation

iterations together. Obtaining the interference waveform due to interferers in both

adjacent- and second-layer cells therefore requires three simulation iterations. Again,

this assumption is technology-independent.

3. All interfering cells transmit on the same RF carrier frequency. Cells not contributing to

the overall interference waveform (e.g., resulting from a cell reuse pattern) may be

“zeroed out” during the computation.

4. Symbols are not aligned from interferer to interferer. There is a random uniform 
 ! 0,T

delay incorporated in each symbol stream.

5. Phase offset is random from interferer to interferer. There is a random uniform 
 ! 0 2, 

constant phase term added to each interferer’s modulated phase. Effectively, this phase

term results in a rotational offset of the OQPSK phase constellation.

A Matlab program was developed for simulation of the CDMA-PCS uplink interference waveform.

Major steps, listed in order of execution, are given below.

1. Load all parameters for the simulation. These parameters, with corresponding

descriptions and values, are given in Table 5.2.

74
Table 5.2. Parameters for CDMA-PCS Uplink Interference Simulation

Parameter/Function Description Value

fc

RF channel carrier frequency; included as a parameter for

frequency-dependent channel models ranges from 1850 to 1990 MHz

Pav Average (maximum) transmit power of each mobile station 2 W

Tchip Chip duration for the I, Q, and long code pseudorandom

sequences

0.813802 x 106 s

Nchips Total number of chips used in the simulation; determines

the length of the interference waveform 2000

Ng Number of samples/phase change 4

Nu

Number of active mobile stations per RF carrier in the

primary cell 20

NuI

Number of active mobile stations per RF carrier per

interfering cell 20

c Interference waveform propagation speed 3 x 108 m/s

R Cell radius 500 m

dxk

Linear X distance of each interfering cell’s base station

with respect to the primary cell’s base station

1000 (depends on R and the

geographical location of the

interfering cell)

d yk

Linear Y distance of each interfering cell’s base station

with respect to the primary cell’s base station

1000 (depends on R and the

geographical location of the

interfering cell)

Nic Number of interfering cells

Varies according to the iteration;

equals 6 for adjacent cells, but

may be set to any number

a r kn kn kn ,, , 
  , Propagation loss/Doppler shift function; may also be set

explicitly to demonstrate a particular power distribution

Default is a simple 1/r4

propagation loss law

2. Design a baseband filter that is CDMA-PCS emissions-compliant. Baseband filtering

tolerances are specified in terms of a minimum allowed mean squared error in document

JTC(AIR)/94.11.21-022R8. Any filter that matches the specifications may be used. The

ITS N/I model implements a Kaiser FIR filter design based on stopband level and

frequency, passband frequency, and allowed passband ripple. Default parameters are

summarized in Table 5.3.

Table 5.3. CDMA-PCS Filter Parameters

Parameter/Function Description Value

fp Passband cutoff frequency 590 kHz

fst Stopband cutoff frequency 740 kHz

Ap Allowed passband ripple 3 dB

As Stopband level, relative to the passband 40 dB

75
3. Calculate the positions of users in the primary cell. Mobile stations are randomly placed

in the primary cell. The base station is assumed to be at the origin. Default distribution

of mobile stations is random uniform. In polar coordinates, this means that the angular

distribution isU ,  0 2 , and the distance rfrom the base station has pdf 2

2

r

R , where R is the

cell radius. The coordinates for each mobile station are stored in polar form.

4. Calculate the propagation loss/Doppler shift function for each mobile station in the

primary cell. Any propagation loss/Doppler shift function may be implemented. The

default is a simple 1

4 r propagation loss law.

5. Load all required spreading codes. Spreading codes are computed offline using a separate

program. For this simulation, only the I and Q spreading codes were explicitly generated

and loaded (long codes are too memory-intensive to simulate explicitly). Orthogonal

modulation and long code spreading is simulated using a random number generator.

6. Set the power profile for primary cell mobile stations. The amplitude weighting

function W t k n, ( ) given in Equation (5.13) may be explicitly set to match the power

received from each interfering mobile station in the primary cell. This method was used

in the ITS N/I model for different interferer amplitude distributions. Alternatively, the

functions A t k n, ( ) and a r kn kn kn ,, , 
  , may be set to reflect the power control algorithm

and propagation loss function, respectively.

Steps 7-9 are executed once for every mobile station in the primary cell.

7. Spread the data stream for the n

th mobile station in the primary cell. The data stream for

the n

th mobile station in the primary cell is created using a random number generator,

split into I and Q streams, and modulo-2 added to the preloaded I or Q codes. All mobile

stations within the primary cell use identical I and Q code offset values. Zero offset is

assumed for I and Q codes within the primary cell.

8. Filter and OQPSK modulate the n

th mobile station’s symbol stream. After spreading, the I

and Q symbol streams are baseband filtered (using the filter designed in step 2). The Q

symbol stream is offset by one-half-symbol duration, then quadrature modulated. A random

U , 
 ! 0 2 phase and randomU ,T 
 ! 0 delay is incorporated as part of the calculation.

9. Add the n

th mobile station’s interference waveform to the aggregate waveform. This is

performed iteratively for every mobile station in the primary cell.

Next, the interference waveform components from mobile stations in the k

th interfering

cell are calculated. Steps 10-12 are executed for each interfering cell.

76
10. Calculate positions of users in the k

th interfering cell. Mobile stations are randomly

placed in the k

th interfering cell. The kth base station is assumed to be at the origin.

Default distribution of mobile stations is random uniform. In polar coordinates, this

means that the angular distribution isU ,  0 2 , and the distance from the base station r

has pdf 2

2

r

R . The coordinates for each mobile station are stored in polar form.

11. Calculate propagation loss/Doppler shift function for each mobile station in the k

th

interfering cell. Any propagation loss/Doppler shift function may be implemented here.

The default is a simple 1

4 r propagation loss law.

12. Set the power profile for k

th interfering cell mobile stations. The functions A t k n, ( ) and

a r kn kn kn ,, , 
  , are set to reflect the power control algorithm and propagation loss

function, respectively, with respect to the n

th mobile station in the kth interfering cell.

The product function W t k n, ( ) is not set explicitly because power control does not exist

between the k

th interfering cell and the primary cell.

Steps 13 -15 are computed for each mobile station in the k

th interfering cell.

13. Spread the data stream for the n

th mobile station. The data stream for the nth mobile

station in the k

th interfering cell is created using a random number generator, split into I

and Q streams, and modulo-2 added to the preloaded I or Q codes. The I and Q codes are

offset by at least 3*64 chip durations, relative to the I and Q codes of every other cell

(including the primary cell) in the simulation. All mobile stations within the k

th

interfering cell use identical I and Q code offset values.

14. Filter and OQPSK modulate the n

th mobile station’s symbol stream. After spreading, the I

and Q symbol streams are baseband filtered (using the filter designed in step 2). The Q

symbol stream is offset by one-half symbol duration, then quadrature modulated. Random

U , 
 ! 0 2 phase and randomU ,T 
 ! 0 delays are incorporated as part of the calculation.

15. Add the n

th mobile station’s interference waveform to the aggregate waveform. This is

completed iteratively for every mobile station in the n

th interfering cell.

16. Compute the interference waveform statistics. These include spectral plots, voltage

envelope histograms, power envelope histograms, and phase distributions.

5.7.2 Example Results

As was the case for the PCS 1900 example, many possible scenarios may be investigated by

choosing the parameters of the CDMA-PCS uplink interference simulation appropriately. For

77
demonstration purposes, three cases were developed and simulated using the parameters specified

in Tables 5.2 and 5.3.

Case 1: Twenty mutually interfering mobile stations in the primary cell only, assuming

perfect power control.

Case 2: Twenty mutually interfering mobile stations in the primary cell only, perfect power

control for all but one mobile station, received 10 dB above the others.

Case 3: Aggregate interference waveform from 6 adjacent cells, with 20 interfering mobile

stations in each adjacent cell.

All cases were simulated for 2000 chip durations. The results are summarized below.

Results for Case 1. Case 1 demonstrates the interference waveform generated by 20 mobile

stations on an uplink RF channel, without any intercell interference. Perfect power control assumes

all mobile station transmissions are received at the base station at equal power levels. As a result,

the default propagation loss law is not implemented, since the receive level at the base station is set

explicitly. All interferer levels were normalized before the statistical analysis was performed.

Figure 5.12 shows the voltage envelope histogram of the interference waveform incident at the

primary base station’s receiver. As was expected, the envelope appears to have a Rayleigh

distribution. The corresponding power envelope histogram shown in Figure 5.13 is approximately

exponentially distributed, and the phase envelope histogram plotted in Figure 5.14 appears to have

a uniform distribution. Hence, the aggregate waveform due to uplink interferers within the primary

cell appears as a Gaussian process. However, reception of a logical channel also depends on the

correlative properties of the individual spreading codes, which are included in the interference

waveform simulation, but not discernible in the histogram plots.

Results for Case 2. In case 2, the same parameters were used as in case 1, except one mobile station

waveform is received at a power level 10 dB above the others. This is performed to illustrate a

severe case of power control imbalance. An actual scenario may demonstrate a more variable

distribution of mobile station power levels received at the base station.

Figure 5.15 shows the voltage envelope histogram of the interference waveform incident at the

primary base station’s receiver. This time, the envelope appears shifted to the right. The

corresponding power envelope histogram shown in Figure 5.16 no longer appears exponentially

distributed, and the phase envelope histogram plotted in Figure 5.17 now contains four clearly

defined states. These states correspond to four randomly shifted phase states from the dominant

78
mobile station’s OQPSK modulated waveform (the fourth state is “wrapped” around the # phase

value). As a result, the aggregate waveform due to uplink interferers within the primary cell is not a

Gaussian process. Although a 10-dB fluctuation may be unrealistic in an actual deployment, more

subtle variations in the power control algorithm may also have a drastic impact on system capacity.

For this reason, the N/I model presented here allows any distribution of interferers to be included in

the aggregate waveform.

Results for Case 3. This case demonstrates the statistical properties of the aggregate interference

waveform from nearby cells. Six interfering cells containing 20 mobile stations each were

simulated. The aggregate interference waveform does not include transmissions from mobile

stations in the primary cell. Since power control is not coordinated between the primary base station

and mobile stations in nearby cells, the default propagation loss law was assumed, with all mobile

stations transmitting at equal power. This assumption may not be realistic, since mobile stations at

the border of their assigned cells will typically transmit at a higher level. In addition, a number of

these mobile stations will tend to be much closer to the primary cell than others.

Figures 5.18, 5.19, and 5.20 show the voltage envelope, power envelope, and phase histograms,

respectively. Scales in this case are not normalized, but individual power levels are accurate with

respect to each other. Statistically, the aggregate waveform appears to have a Gaussian distribution.

Correlative codes incorporated into the aggregate waveform may have additional impacts on the

actual performance. For demonstration purposes, 100 mobile stations in the primary cell were

simulated. The resulting interference spectrum is shown in Figure 5.21. The simulation program is

capable of simulating an arbitrary number of mobile stations in a given cell.

79
80

Figure 5.12. Voltage envelope histogram for case 1.

Figure 5.13. Power envelope histogram for case 1.
81

Figure 5.14. Phase envelope histogram for case 1.

Figure 5.15. Voltage envelope histogram for case 2.
82

Figure 5.16. Power envelope histogram for case 2.

Figure 5.17. Phase envelope histogram for case 2.
83

Figure 5.18. Voltage envelope histogram for case 3.

Figure 5.19. Power envelope histogram for case 3.
84

Figure 5.20. Phase envelope histogram for case 3.

Figure 5.21. Aggregate interference waveform spectrum caused by 100 mobile stations in the

primary cell.
6. SUMMARY AND CONCLUSIONS

A generic methodology for modeling noise and interference for cellular wireless technologies was

presented in this report. Specifically, the Institute for Telecommunication Sciences has developed

models of the self-interference created by a PCS network of individual communication links. This

generic methodology was then applied to two proposed PCS systems: GSM-based PCS 1900 and

IS-95-based CDMA.

Accurate noise and interference models are critical for meaningful PCS network testing, modeling,

and evaluation. Advanced wireless systems can be tested via 1) channel simulation in real time or 2)

large-scale software simulation for performance and interoperability evaluation. Noise and

interference models described in this report are suitable for both applications. These capabilities

directly benefit other Federal agencies with special performance requirements for PCS services;

they also benefit small equipment and service providers without the resources to evaluate PCS

technologies and equipment.

Using a geometric framework, generic expressions were developed for both uplink and downlink

interference. The generic expressions then were modified for application to specific PCS systems.

Models for PCS 1900 and CDMA-PCS were developed by extracting elements of each system that

affect the noise and interference characteristics, and applying them to the modified generic

expressions. Resulting expressions were specific to the PCS technology. Working examples of

models for PCS 1900 and CDMA-PCS were then presented. In addition, computer simulation of PCS

1900 and CDMA-PCS was used to produce and analyze sample noise and interference waveforms.

Models of the noise and interference are essential for realistic characterization of the operating

environment for proposed PCS technologies. Examples for both PCS 1900 and CDMA-PCS

demonstrated that many enviromental and systematic influences affect the statistics of the aggregate

interference waveform, and that a Gaussian distribution assumption is too simplistic in many cases.

In a future report, efficient real-time implementation of the PCS system-specific noise and

interference waveforms will be described. Network-level simulations for interoperability and

performance evaluation of proposed wireless technologies, which make use of the noise and

interference models, will be presented in another future report.

85

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Symposium on Personal, Indoor, and Mobile Radio, P 3.2, pp. 1280-1284, 1994.

[10] R. Prasad and A. Kegel, “Improved assessment of interference limits in cellular radio

performance,” IEEE Trans. Vehic. Tech., Vol. 40, No. 2, pp. 412-419, May 1991.

[11] J. Zou, V.K. Bhargava, and Q. Wang, “Reverse link analysis and performance

evaluation for DS-CDMA cellular systems,” in Proc. IEEE International Symposium

on Personal, Indoor, and Mobile Radio, BC 1.2, pp. 50-54, 1994.

[12] R.F. Diesta and J.P. Linnartz, “Using Laplace transforms to compute performance of

mobile radio links,” in Proc. IEEE International Symposium on Personal, Indoor, and

Mobile Radio, B 3.3, pp. 301-305, 1994.

87
[13] B.D. Woerner, J.H. Reed, and T.S. Rappaport, “Simulation issues for future wireless

modems,” IEEE Communications Magazine, Vol. 32, No. 7, pp. 42-53, Jul. 1994.

[14] A.G. Burr, T.C. Tozer, and S.J. Baines, “Capacity of an adaptive TDMA cellular

system: comparison with conventional access schemes,”in Proc. IEEE International

Symposium on Personal, Indoor, and Mobile Radio, E 2.2, pp. 242-246, 1994.

[15] Z.J. Haas, J.H. Winters, and D.S. Johnson, “Simulation study of the capacity bounds in

cellular systems,” in Proc. IEEE International Symposium on Personal, Indoor, and

Mobile Radio, B 7.2, pp. 1114-1120, 1994.

[16] M.A. Couture and J.P. Linnartz, “Improved channel modeling for performance analysis

of a DS-CDMA link,” in Proc. IEEE International Symposium on Personal, Indoor,

and Mobile Radio, C 8.3, pp. 705-710, 1994.

[17] Y. Li, B.D. Woerner, W Tanis II, and M. Hughes, “Simulation of CDMA using

measured channel impulse response data,” in Proc. IEEE Vehic. Tech. Conf., pp.

428-431, 1993.

[18] J.G. Ferranto and J.J. Lemmon, “Simulation model of interference in PCS frequency

bands,” JTC(AIR)/93.08.23, Aug. 23, 1993.

[19] Personal Communications Services Air Interface Specification (ANSI

J-STD-007-1996) (T1/TIA Joint Standard), American National Standards Institute, 11

West 42nd Street, New York, NY 10036, 1996.

[20] J.J. Tuma, Handbook of Numerical Calculations in Engineering, New York:

McGraw-Hill, Inc., 1989, p. 196.

[21] “Draft Proposed American National Standard - Personal Station - Base Station

Compatibility Requirements for 1.8- to 2.0-GHz Code Division Multiple Access

(CDMA) Personal Communications Systems” T1 J-STD-008, Dec. 12, 1994.

88
APPENDIX A: MODULATED WAVEFORM NOTATION

This appendix describes the notation used for continuous modulated waveforms throughout this

document, and contains a brief introduction to the theory of phase modulation.

A generalized modulated waveform can be expressed in the form:

st At ft t ! c ( , ) ( , )cos ( , )    
 2  (A.1)

where represents values of a digital information alphabet sequence. Information can be included

in the waveform by modulating the phase, rate of change of the phase (frequency), amplitude, or a

combination of all the above. For a constant amplitude phase-modulated waveform, only(, ) t  is a

function of the digital information alphabet sequence. Letting

A t E

T

s (, )   2 (A.2)

yields the phase-modulated waveform:

s t ! E

T ft t s

c ( , ) cos ( , )    


2 2  (A.3)

where Es is defined as the energy per symbol, and T is the symbol duration. The phase expression

(, ) t  can be expressed in terms of the modulation index h, individual values of the digital

information alphabet sequence  i

, and an arbitrary pulse q t( ):

  ( , ) ( ). t h q t iT i

i

  



2  (A.4)

For a general waveform xt t ( ) cos ( )   !, the instantaneous frequency is defined as:

f t d t

dt I ( ) ( )  12

 (A.5)

Then, for the total angle expression   () (, ) t ft t  
 2 c  in Equation (A.3), the instantaneous

frequency expression becomes, using Equations (A.4) and (A.5):

f t f h q t iT

f h

I c i

i

c i

i

() ( )  
 + 











 








 1

2

2 2



 













g t iT ( ), where 

q(t) = g( )d .

=-

$ $

$

(A.6)

89
Therefore, the pulse q t( )modulates the phase of the waveform s t(, )  , and its derivative g t( )defines

the instantaneous frequency. The function g t( )may be chosen from numerous pulse shapes, and its

shape determines the smoothness of the phase. Proper selection of g t( ) results in a PCS RF

waveform that utilizes bandwidth efficiently.

90
APPENDIX B: COMPLEX BASEBAND NOTATION

This appendix is adapted from [1], and describes the theory and notation used for handling

bandpass waveforms in complex baseband form. This is necessary because actual carrier

frequencies used in proposed PCS systems are much greater than the waveform bandwidth,

resulting in a prohibitively large Nyquist sampling rate.

The pre-envelope of a real bandpass waveform g t( ) is defined as:

g t g t jg t 
 () ()  
 ( ), (B.1)

where g t ( )is the Hilbert transform of g t( ). The Hilbert transform of the waveform g t( )is expressed as:

() () ,

( ) ( )( sgn( )).

gt gt

t

Gf Gf j f

 

 

1



or (B.2)

Note that G f 
 ( )is equal to G f ( )for frequencies greater than zero (except for a scalar multiplier),

and equals zero for all frequencies less than zero.

The pre-envelope g t 
 ( ) can also be expressed in terms of a complex envelope g t( ), which forms a

complex baseband representation of the real-valued bandpass waveform g t( ). g t( ) is used for

practical considerations; the sampling rate required at the RF carrier frequency for PCS systems is

prohibitively large compared to the bandwidth of the waveform. Using complex envelope notation

allows complete representation of the waveform within the complex baseband bandwidth, which is

equal to the bandpass bandwidth of the actual (real) waveform.

The pre-envelope may be expressed in complex exponential form as follows:

g t gt j ft 
  c ( ) ( )exp( ). 2 (B.3)

In Equation (B.3), g t( ) is a real-valued bandpass waveform at carrier frequency f c and g t( ) is the

complex baseband representation of the waveform g t( ). Then

g t jg t g t j f t

gt ft j

c

c

( ) ( ) ( )exp( )

( ) cos( ) sin(


 

 


2

2 2



 !  f t c ) .

(B.4)

Because g t( ) is generally complex,

g t g t jg t c s () () ()  
 (B.5)

91
where g t c ( )and g t s( )are both real-valued functions. Using Equations (B.1), (B.3), and (B.5), g t( )is

expressed in canonical form as:

gt g t g t ft g t ft ! c cs c ( ) Re ( ) ( )cos( ) ( )sin( ).  
 2 2   (B.6)

g t c ( )and g t s( )are known as the in phase and quadrature components of the waveform g t( ). Both are

lowpass waveforms.

Filter implementations and transfer functions also may be formulated in complex baseband form.

Below it is demonstrated that the real filter equation yt xt ht () () ()   may be replaced by the

equivalent complex baseband equation yt xt ht () () ()   . To begin, a bandpass waveform x t( ) and

a transfer function h t( ) are defined in canonical form:

xt x t ft x t ft c cs c ( ) ( )cos( ) ( )sin( )  2 2   (B.7)

ht h t f t h t f t c cs c ( ) ( )cos( ) ( )sin( ).  2 22 2   (B.8)

The constant “2” in Equation (B.8) is arbitrary. The waveform x t( )is input to a filter with transfer

function h t( ). Using the complex baseband representation for h t( ):

ht ht j f t ! c ( ) Re ( )exp( ) .  2 2 (B.9)

Next, the waveform at the output port of the filter h(t) is defined as:

yt yt j ft h xt d ! c ( ) Re ( )exp( ) ( ) ( ) .  





 2 $ $$ (B.10)

Therefore,

yt h x t d ( ) Re ( ) Re ( ) .  ! ! 
 






2 $ $$

$

(B.11)

To show that yt xt ht () () ()   , an integral identity is needed. Given an expression

m t g t g t dt

t

( ) Re ( ) ( ) *  













 
 






 1

2 1 2 (B.12)

where

g t g t jg t

g t g t jg t

11 1

22 2







 


 


() () ( )

() () ( ),

(

(

(B.13)

92
g t 1 ( ) and g t 2 ( ) are real functions, g t ( ) 1 and g t ( ) 2 are their respective Hilbert transforms, and “*”

denotes complex conjugation. Equation (B.12) then becomes:

m t g t g t jg t g t jg t g t g ( ) Re ( ( ) ( ) ( )  

 ( ) () () 1

2 12 12 12 1 ( )t g t dt ( ))

t

2























(B.14)

Because g t 1 ( ) and g t 2 ( ) are real functions, Equation (B.14) can be reduced to

m t g t g t dt g t g t dt

t t

() () ()  
 ( )() .









  1

2

1

2 12 12 (B.15)

The cross-correlation of two Fourier-transformable functions has cross-power spectral density

equal to the Fourier transform of the first function multiplied by the complex conjugate of the

Fourier transform of the second function. This is given by the correlation theorem as:

FT ! R G fG f j fG fG f G f 12 1 2

2 2

12 1 ( ) ( ) ( ) sgn ( ) ( ) ( ) ( ) * * $   G f 2

*

( ) (B.16)

Taking the inverse Fourier transform of the expression to the right of the last equality yields the

cross-correlation of the Hilbert transforms of g t 1 ( )and g t 2 ( ), respectively. Thus, from Equation (B.15):

1

2

1

2 12 12 g t g t dt g t g t dt

t t

() () ( )( )









   (B.17)

and Equation (B.15) becomes:

m t g t g t dt g t g t dt ! ! t t( ) ( ) ( ) Re ( ) Re ( ) *  






 




 1

2

1

2 12 1 2





 . (B.18)

Returning to Equation (B.12), and using Equation (B.17), the identity becomes

m t g t g t dt g t g ! t( ) Re ( ) ( ) Re ( ) Re *  













 
 
 





 


1

2 12 1 ! 2






 * () . t dt

t

(B.19)

Now, if g t 2 ( ) is replaced with gt gst 3 2 () ( )  , the identity becomes:

m t g t g s t dt g t ! t( ) Re ( ) ( ) Re ( ) R *  













 
 
 





 


1

2 12 1 e() ! * g s t dt

t

2






 (B.20)

which is the desired identity. Using Equation (B.20), Equation (B.11) becomes:

93
yt h x t d ! !

h xt d

( ) Re ( ) Re ( )

Re ( ) ( )

 

 


 







 




2 $ $$

$ $$

$

$ 





















 Re ( )exp( ) ( )exp( ( )) h j ftxt j f t c c $  $ $ 2 2 d

h xt d j ft c

$

$ $$ 

$

$





























 



Re ( ) ( ) exp( ) 2

 !













  Re ( ) ( )exp( ) . ht xt j ft 2 c

(B.21)

Therefore, the complex system output y t( )can be obtained by convolving the complex envelopes of

x t( ) and h t( ), and all bandpass calculations may use an equivalent complex baseband representation

of the real waveforms and transfer functions.

REFERENCE

[1] S. Haykin, Communication Systems, Second Edition, New York: John Wiley and Sons,

Inc., 1983.

94
APPENDIX C: ACRONYMS

C/I carrier-to-interference ratio

C/(N +I) carrier-to-noise plus interference ratio

CDMA code-division multiple access

CDMA-PCS an IS-95-based standard for 1900 MHz PCS

FDD frequency-division duplexing

FWPC Federal Wireless Policy Committee

GMSK Gaussian minimum-shift keying

GSM Global System for Mobile

I inphase waveform component

IS-95 CDMA cellular standard

ITS Institute for Telecommunication Sciences

JTC Joint Technical Committee

LPF lowpass filter

N/I noise/interference

NTIA National Telecommunications and Information Administration

OQPSK offset quadrature phase-shift keying

PACA priority access and channel assignment

PCS personal communications services

PCS 1900 a GSM-based standard for 1900 MHz PCS

PDF probability density function

Q quadrature waveform component

QPSK quadrature phase-shift keying

RF radio frequency

SNR signal-to-noise ratio

STU-III secure telephone unit - III

TDMA time-division multiple access

95