Comparison of Electromagnetic Simulation Results 

with Experimental Data for an Aperture-Coupled 

C-band Patch Antenna 

by Steven Keller, William Coburn, 

Theodore Anthony, and Chad Patterson 

ARL-TR-3994 November 2006 

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Army Research Laboratory 

Adelphi, MD 20783-1197 

ARL-TR-3994 November 2006 

Comparison of Electromagnetic Simulation Results 

with Experimental Data for an Aperture-Coupled 

C-band Patch Antenna 

Steven Keller, William Coburn, Theodore Anthony, 

and Chad Patterson 

Sensors and Electron Devices Directorate, ARL 

Approved for public release; distribution unlimited. 
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Comparison of Electromagnetic Simulation Results with Experimental Data for 

an Aperture-Coupled C-band Patch Antenna 

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Steven Keller, William Coburn, Theodore Anthony, and Chad Patterson 

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U.S. Army Research Laboratory 

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14. ABSTRACT 

The Method of Moments, implemented in 2.5-dimensions (2.5-D) with the multilayer Green’s Function, and the Finite Element Method, a 

fully three-dimensional (3-D) solution of Maxwell’s equations, are two popular methods for antenna simulations. Commercial software 

implementing these two methods was used to simulate a single aperture-coupled patch antenna element. This effort was time limited so that a 

simple antenna was selected for which the antenna design, fabrication, and measurements could be accomplished in less than three months. 

The simulation results are compared with experimental data measured for an antenna prototype that has substrate and ground plane dimensions 

of 4 inches by 4 inches. The microstrip feed line extends to the edge of the substrate where a coaxial connector is installed between the 

microstrip and slotted ground plane and there is no bottom ground plane. The results are used to compare and contrast the use of these two 

approaches for the design and simulation of an aperture-coupled patch antenna with finite size substrate and ground plane. 

15. SUBJECT TERMS

Aperture coupled patch, impedance bandwidth, microstrip, FEKO, EMPiCASSO 

16. SECURITY CLASSIFICATION OF:

19a. NAME OF RESPONSIBLE PERSON

Steven Keller 

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(301) 394-2705 

Standard Form 298 (Rev. 8/98) 

Prescribed by ANSI Std. Z39.18 
 

Contents 

List of Figures iv

List of Tables iv

Acknowledgment v

1. Introduction 1

2. EMPiCASSO Simulation 1

3. HFSS Simulation 3

4. Experimental Data 5

4.1 Network Analyzer Measurements ......................................................................................5 

4.2 Antenna Pattern Measurements..........................................................................................5 

5. Results 8

6. Conclusion 13

7. References 14

Distribution List 15 

 iii
 

List of Figures 

Figure 1. An aperture-coupled patch antenna design in EMPiCASSO showing (a) the 

substrate configuration and (b) the mesh discretization. ...........................................................2 

Figure 2. Aperture-coupled patch antenna model in HFSS showing the 3-D model and the 

radiation boundary. ....................................................................................................................3 

Figure 3. Radiation pattern measurement setup in the anechoic chamber......................................6 

Figure 4. Schematic diagram of antenna measurements with the substitution method...................7 

Figure 5. VSWR comparison for an aperture-coupled C-band patch antenna. ..............................8 

Figure 6. S11 comparison for an aperture-coupled C-band patch antenna. .....................................9 

Figure 7. Measured realized gain compared to EMP directivity and HFSS realized gain for 

an aperture-coupled C-band patch antenna..............................................................................10 

Figure 8. Measured gain compared to EMP and HFSS directivity for an aperture coupled 

C-band patch antenna...............................................................................................................11 

Figure 9. E-Plane radiation pattern comparison at 4.55 GHz.......................................................12 

Figure 10. H-Plane radiation pattern comparison at 4.55 GHz.....................................................12 

List of Tables 

Table 1. Aperture-coupled patch antenna design parameters. ........................................................2 

Table 2. HFSS simulation parameters for an aperture-coupled patch antenna...............................4 

Table 3. Typical parameters obtained for antenna measurements...................................................7 

 iv
 

Acknowledgment 

The authors would like to thank Russell Harris of the Army Research Laboratory (ARL) for 

converting the antenna geometry into fabrication drawings and for constructing the prototype. 

Steven Weiss of ARL provided guidance and advice in the antenna measurements along with a 

critical review of the manuscript. Their assistance on this project was greatly appreciated. 

 v
 

INTENTIONALLY LEFT BLANK. 

 vi
 

1. Introduction 

With the large number of electromagnetic (EM) simulation software packages available for use 

in antenna design, which use different numerical techniques in the time or frequency domain, it 

is often difficult to determine which program will work best for a given antenna geometry. For 

multilayer planar structures, the Method of Moments (MoM) or Finite Element Method are quite 

popular. In order to streamline the antenna design process and generate accurate results before 

prototype construction, it is important to select an EM simulation program that will provide an 

optimal balance between a minimal simulation run time and a maximized correlation between 

the simulation results and the experimental data. An aperture-coupled C-band patch antenna is 

designed and simulated with both EMAG Technologies EMPiCASSO (EMP) (1) and Ansoft’s

High Frequency Structure Simulator (HFSS) (2). These codes were chosen as representative of 

the MoM implemented in 2.5-dimensions (2.5-D) with the multilayer Green’s Function (EMP) 

and the FEM, a fully three-dimensional (3-D) solution of Maxwell’s equations (HFSS). These 

simulation results are compared with experimental data measured for an antenna prototype that 

has substrate and ground plane dimensions 4 inches by 4 inches. The feed line extends to the 

edge of the substrate where a coaxial connector is installed between the feed and slotted ground 

plane and there is no bottom ground plane. The results are used to compare and contrast the use 

of EMP and HFSS for the design and simulation of an aperture-coupled patch antenna with finite 

size substrate and ground plane. This effort was time limited so that a simple patch design was 

chosen for which the antenna design, fabrication, and measurements could be accomplished in 

less than three months. 

2. EMPiCASSO Simulation 

A C-band patch antenna was designed as a 2.5-D model in EMP to operate with a resonant 

frequency between 4.4 and 5 GHz. Since the patch antenna is inherently a narrowband structure, 

it was not expected to cover this entire band, so the design focused on the center frequency 

around 4.6 GHz corresponding to a free-space wavelength, λ0 = 2.57 inches. The goal in the 

antenna design was to position the center frequency around 4.6 GHz and maximize the –10-dB 

return loss bandwidth. The final antenna design is shown in figure 1 with the detailed 

dimensions and substrate layer properties listed in table 1. The relative dielectric constant of the 

substrate layers were set to εr = 2.33 to represent Rogers RT/duroid1 5870 low-loss dielectric (3). 

Then the effective wavelength in the substrate, λeff = λ0/√εr = 1.68 inches is used for mesh 

refinement in the aperture and on the microstrip feed with 80 samples/λeff rather than the default 

 1

 Duroid is a registered trademark of Rogers Corporation. 

 1
 

 2

30 samples. The 5870 substrate is 2-oz copper clad amounting to about 70 μm copper thickness, 

and the different layers are bonded with 3M Scotch2

 adhesive transfer tape3. 

(a) 

(b) 

Figure 1. An aperture-coupled patch antenna design in EMPiCASSO showing (a) the substrate configuration and 

(b) the mesh discretization. 

Table 1. Aperture-coupled patch antenna design parameters. 

Layer Description Dimensions and Substrate Thickness (mil) 

Patch (Copper) 680- by 860-mil patch 

Upper Substrate (RT/duroid) 4000 by 4000 mils, 125-mil thickness 

Slotted Ground Plane (Copper) 4000 by 4000 mils with 50- by 430-mil slot 

Lower Substrate (RT/duroid) 4000 by 4000 mils, 20-mil thickness 

Microstrip Feed Line (Copper) 

55-mil width, 

2200-mil total length 

extending 200 mils past slot center 

 2

 3M and Scotch are registered trademarks of 3M Corporation. 

3

 http://solutions.3m.com/en_US/. 
 

The design involved a number of modeling approximations, including an infinite slotted ground 

plane and infinite substrate, with a feed line trace that EMP automatically extends to ~2λeff at 

each excitation frequency. A port excitation was placed along this feedline roughly at the center 

and simulated with the MoM technique in EMP. A convergence study showed that the default 

mesh size was sufficient except for the slot and microstrip feed line where mesh refinement is 

needed to improve accuracy. The simulation took ~30 seconds per frequency or about 10 

minutes to complete a 21-point sweep from 4.4 to 4.8 GHz. The 2.5-D simulation results are 

compared to the experimental data for a prototype antenna fabricated on a 4-inch-square 

substrate. 

3. HFSS Simulation 

The aperture-coupled patch antenna that was designed and fine tuned in EMP was then 

constructed as a 3-D model in HFSS. This model, along with the radiation boundary, is shown in 

figure 2. The substrate dimensions were set to the as-fabricated antenna dimensions, 4000 by 

4000 mils where the feed line extends 2000 mils from the aperture center to the ground plane 

edge. A coaxial excitation was applied at the end of the feed line rather than a lumped port 

excitation as in EMP. A radiation boundary was constructed around the model which extended 

1500 mils from each edge of the substrate corresponding to ~0.58λ0. 

Figure 2. Aperture-coupled patch antenna model in HFSS 

showing the 3-D model and the radiation boundary. 

 3
 

A number of different simulations were run with this model, with certain simulation parameters 

such as the max delta S and the lambda refinement target being altered before each trial run. The 

max delta S parameter is the criterion for convergence, with the simulation continuing to refine 

its adaptive mesh until the S-parameters of the simulated solution between two consecutive 

iterations differ by less than the value of this parameter. Thus, with a smaller max delta S, a 

stronger convergence criterion will be satisfied, but it will take more adaptive passes to converge 

and the simulation time will consequently increase. The lambda refinement target parameter sets 

the approximate size, in wavelengths, of each cell on the adaptive mesh that is created over the 

simulated structure. Thus, with a smaller lambda refinement target, the adaptive mesh will 

contain more elements, potentially offering a more accurate solution at the cost of increased 

simulation run time and increased system memory usage. The details of these simulations are 

shown in table 2 for a 21-point frequency sweep (4.4 to 4.8 GHz) with 20% maximum mesh 

refinement per pass. 

For each of these parameter choices, a fast and discrete sweep was run to compare simulation 

sweep methods. The fast-sweep simulation has a much shorter simulation run time than the 

discrete-sweep simulation, which solves each frequency point separately but consumes far more 

system memory. For a small frequency range such as used in these simulations, the fast-sweep 

simulation is ideal from an efficiency standpoint and similar to the EMP run time. The 

simulation 4 results are used for comparison with the EMP results and with experimental data 

measured for an antenna prototype. 

Table 2. HFSS simulation parameters for an aperture-coupled patch antenna. 

Simulation Parameters Simulation Data 

Simulation 

Max 

Delta 

S 

Lambda 

Refinement 

Target 

Tetrahedra in 

Mesh 

Passes before 

Convergence 

Run time 

(min.) 

|S11| 

Minimum 

(dB) 

1 – Fast 0.02 0.3333 11841 10 5.5 –32.7 

1 – Discrete 0.02 0.3333 14236 11 30 –33.3 

2 – Fast 0.01 0.3333 14158 11 6 –36.7 

2 – Discrete 0.01 0.3333 17085 12 21 –33.1 

3 – Fast 0.02 0.25 18753 9 8 –37.2 

3 – Discrete 0.02 0.25 15646 8 19.5 –44.4 

4 – Fast 0.01 0.25 18795 9 8.5 –45.4 

4 – Discrete 0.01 0.25 27075 11 39 –28.9 

 4
 

4. Experimental Data 

A prototype of this aperture-coupled patch antenna design was constructed with an on-site router 

on RT/duroid substrates, which are then adhesive bonded. The 5870 has a relative dielectric 

constant of approximately εr = 2.33 as measured with the split cavity method by Damaskos, Inc.4 

The return loss data were measured with a network analyzer, and the gain and radiation patterns 

were measured in an on-site tapered anechoic chamber. 

4.1 Network Analyzer Measurements 

The dielectric and conductor losses are calculated in HFSS as independent loss mechanisms, but 

for typical materials, these losses are negligible. The impedance mismatch to the antenna 

determines the input reflection coefficient, Γ, or in terms of S-parameters, Γ = S11. Performance 

is often characterized by the return loss, RL = 20log(|Γ|), or alternatively by the input voltage 

standing wave ratio, VSWR = (1 + |Γ|)/(1 − |Γ|). Typical C-Band specifications are VSWR < 1.6 

over the range 4.4 to 5 GHz corresponding to RL < −13 dB. The measured data were obtained 

with a Wiltron5 37269A vector network analyzer (VNA) calibrated with the Wiltron K-Cal Kit 

Model 3652. When care is taken to make consistent connections, the measurement repeatability 

is on the order of ±0.05 dB which is more than sufficient for research purposes. 

4.2 Antenna Pattern Measurements 

The radiation pattern and gain measurements were conducted with a C-band standard gain horn 

(SGH) antenna as the system transmitter and the prototype patch antenna set up on a nonmetallic rotating positioner to serve as the receiver. A picture of this procedure is shown in 

figure 3. 

 4

 Damaskos, Inc., Concordville, PA http://www.damaskosinc.com/. 

5

 Wiltron Company, Morgan Hill, CA. 

 5
 

Figure 3. Radiation pattern measurement setup in the anechoic chamber. 

We conducted the antenna measurements in an absorber-lined tapered anechoic chamber (4). 

The length of this chamber is about 50 ft, with the actual distance between transmission and 

reception antennas being 45.3 ft. Typical component values are shown in table 3 where the 

equipment is the same between the reference and test setup. Notice that the measured path loss 

minus the cable and connector attenuation is somewhat less than would be predicted in free 

space (–64.8 dB). This is a well-known artifact of the guided mode structure in tapered anechoic 

chambers that is neglected for the purpose of demonstrating antenna performance. A diagram 

for the antenna pattern measurements is shown in figure 4 with the substitution technique with 

Narda6 SGH antennas. The receiver is phase locked to the transmitter with computer-controlled 

data acquisition and rotation. The pattern measurements are calibrated according to power meter 

data versus frequency at the test antenna boresight position. Magnitude and phase data are 

collected every 1.02 degrees with an accuracy of 0.1 degree, although this does not account for 

position error in placement of the antenna on the rotator (4). 

 6

 http://www.nardamicrowave.com/east/products.cfm. 

 6
 

Table 3. Typical parameters obtained for antenna measurements. 

Received with SGH (R1) Received with Test Antenna (R2) 

Source (SRC) Output Typically +6 dBm Typically +6 dBm 

Cable 1 Attenuation −1 dB −1 dB 

Transmit Antenna (Tx) Gain 15.9 dBi 15.9 dBi 

Path Loss −66.7 dB −66.7 dB 

Receive Antenna (Rx) Gain 15.9 dBi X 

Cable 2 Attenuation −3 dB −3 dB 

Receiver Attenuation 0 dB 0 dB 

Received Power R1 ~ −29 dBm R2 (dBm) 

Correction Factor, Δ(dB) Δ = R2 − R1 ± ΔSRC

Test Antenna Gain (dBi) G = 15.9 + Δ

Tx 

SRC 

R1 or R2

Absorber Lined Tapered Anechoic Chamber 

Cable 1 

RCVR 

Cable 2 45.3 ft

Figure 4. Schematic diagram of antenna measurements with the substitution method. 

 7
 

Two identical Narda SGH antennas that have known gain relative to an isotropic radiator (dBi) 

over the rated bandwidth are used as the reference antennas. The reception SGH is then replaced 

with the antenna undergoing test. The exact position of the reference and test antennas on the 

pylon is a source of uncertainty that is minimized but not completely eliminated. The reception 

pattern for the radiating antenna is measured versus angle with the experimental error estimated 

from the repeatability of the data after the antenna is repositioned. The repeatability error can be 

minimized with careful procedures and placement of the antenna on the rotating pylon but it is 

not negligible, typically ±0.25 dB. 

5. Results 

After the prototype return loss and radiation pattern were measured, the data was then processed 

and compared with the results from the EMP and HFSS simulations. The VSWR comparison is 

shown in figure 5 and the corresponding S11 comparison is shown in figure 6 for measurements 

in the range of 4.4 to 4.8 GHz. The EMP simulation results are very close to measurements and 

this software has been used successfully many times for optimizing an antenna design so that the 

resulting performance meets the antenna specifications. The HFSS result is shifted −90 MHz to 

a resonant frequency, fr = 4.49 GHz, so this simulation would suggest a 20-mil smaller patch 

length (~3% reduction) to obtain the measured resonance at fr = 4.58 GHz. 

Figure 5. VSWR comparison for an aperture-coupled C-band patch antenna. 

 8
 

Figure 6. S11 comparison for an aperture-coupled C-band patch antenna. 

From these plots, it can be seen that the EMP simulation data agree the best with the 

experimental results. While the –10-dB return loss bandwidth of the EMP simulation is 

somewhat larger (330 MHz) than the actual measured bandwidth (300 MHz) of the antenna 

prototype, the VSWR data are in reasonable agreement between 4.4 GHz and 4.8 GHz, and the 

simulated center frequency of ~4.580 GHz is very close to the measured value of ~4.583 GHz. 

The HFSS simulation data seem to be moderately consistent in their results, regardless of the 

simulation parameter values discussed in section 3. The major difference between simulation 

trials is the return loss value at the resonant frequency. While the discrete sweep simulation 4 

resulted in a resonant return loss close to that of the experimentally measured value of −27 dB, 

the rest of the simulations produced more idealized results between −32 and −45 dB. An 

important detracting note is that all the HFSS simulation results appear shifted in frequency from 

the EMP and the experimental results by −90 MHz, with all the simulations displaying resonance 

at ~4.49 GHz. If the HFSS simulation result were shifted 90 MHz, it would be in reasonable 

agreement with the measured data. The explicit reason for this frequency-shifted aberation is 

unknown, since the HFSS model dimensions and simulation parameters were exactly the same as 

the as-fabricated antenna. The difference between EMP and HFSS should only be attributable to 

the 2.5-D versus a 3-D model of the substrates, and this difference is known to be small and 

normally neglected. 

The measured radiation pattern data are compared to simulated radiation patterns from EMP and 

HFSS. The measured data correspond to realized gain, Gr, but only directivity is available from 

 9
 

the EMP 2.5-D simulation. Since the antenna losses are small, the EMP directivity should be 

nearly the same as the gain while the realized gain includes impedance mismatch losses. The 

directivity is assumed to be equivalent to gain, G0; then it can be converted to Gr based on the 

calculated reflection coefficient versus frequency, Gr = G0*(1 – |Γ|

2

). HFSS provides directivity, 

gain, and realized gain, but in these simulations, the observables did not have consistent behavior 

at all frequencies, which implies that our model is not sufficiently refined to provide accurate 

radiation calculations. The measured Gr is compared to the EMP and the HFSS directivity and 

realized gain in figure 7 where the HFSS result is significantly lower than expected. 

Figure 7. Measured realized gain compared to EMP directivity and HFSS realized 

gain for an aperture-coupled C-band patch antenna. 

The measured Gr is converted to measured gain, G0, according to the measured S11 data versus 

frequency, G0 = Gr/(1 - |Γ|

2

). A comparison of the measured gain, the EMP and the HFSS gain 

for a frequency range of 4.4 to 4.8 GHz is shown in figure 8. The G0 obtained from the 

measured Gr near the resonant frequency is about 8 dBi, with EMP being in fair agreement at 

7.4 dBi, but the HFSS result is significantly lower than measured: only 5.3 dBi. The EMP 

directivity was in reasonable agreement with measurements whereas the HFSS gain was ~1.5 to 

3 dB less than measured. A more detailed investigation of this discrepancy is ongoing where 

parameter studies of the radiation boundary and layer details are used to further explore the 

model parameters that most influence the directivity calculated by HFSS. 

 10
 

Figure 8. Measured gain compared to EMP and HFSS directivity for an aperture 

coupled C-band patch antenna. 

The measured and simulated radiation patterns at 4.55 GHz are compared in figure 9 for the Eplane pattern and in figure 10 for the H-plane pattern as a function of elevation angle at fixed 

azimuth. The HFSS simulation results seemed to match the pattern shape of the measured data 

better than the EMP simulation but seemed to be less accurate for the antenna directivity. EMP 

had a more idealized pattern and did not predict the reduced gain on boresight observed in the Eplane pattern in the measured and HFSS results. The EMP simulation results tended to match 

the main lobe reasonably well but did not accurately predict the variations in the back plane 

radiation. The EMP 2.5-D model has backplane radiation because of leakage through the slotted 

ground plane rather than the effects of a finite size ground, so these simulation results are as 

expected. For the E-plane pattern, HFSS has back lobes associated with diffraction effects from 

the finite ground, but it does not accurately account for backward radiation through the slot. 

HFSS obtains similar back lobes in the E-plane pattern as measured but for the H-plane has a 

more idealized back plane pattern. Both methods underestimate the peak back plane radiation 

level by several decibels. 

 11
 

Figure 9. E-Plane radiation pattern comparison at 4.55 GHz. 

Figure 10. H-Plane radiation pattern comparison at 4.55 GHz. 

 12
 

6. Conclusion 

The accuracy and efficiency of the EMAG EMPiCASSO and Ansoft HFSS software packages 

were compared with experimental results for an aperture-coupled C-band patch antenna 

geometry. Although the EMP software employed more approximations in its 2.5-D modeling 

format, it produced far more accurate S-parameter simulation results when compared to 

measured data than did the HFSS 3-D simulation. The EMP simulation time for a 21-point 

frequency sweep is on the same order or faster than any of the HFSS simulations performed in 

this study. While HFSS is widely used to model a variety of antenna structures and provides a 

moderately accurate solution for this aperture-coupled patch antenna, EMP seems to provide 

more accurate and more efficient S-parameter solutions. For radiation patterns, the results 

suggest that neither code will result in an accurate prediction of the true radiation pattern. EMP 

has reasonable accuracy for simulating front plane radiation, while HFSS may be more useful for 

simulating back plane radiation. However, in this study neither approach was sufficient for 

accurately predicting the back plane radiation amplitude or pattern. 

The software explored in this paper offers advantages and disadvantages, depending on the 

parameters that must be simulated. For the antenna input impedance, EMP will offer the most 

accurate and efficient solution. For gain data, neither program will offer fully accurate results, 

but EMP appears to be more accurate compared to the measured data. It was expected that the 

HFSS 3-D model would provide more accurate results for back plane radiation patterns and 

would be useful for exploring the effects of a finite ground plane. It is not clear why the HFSS 

model produced such poor radiation calculations so further investigation is required. The results 

presented may be more sensitive to the size and shape of the radiation boundary than observed to 

date so that more parameter studies are required. Increasing the radiation boundary and refining 

the mesh further requires significantly more computation time so that even if better accuracy 

could be obtained, the simulations would be highly inefficient compared to the speed of EMP. 

For radiation pattern data, EMP provides a sufficiently accurate depiction of the front plane 

pattern with an idealized back plane pattern. HFSS may be useful for finite size planar antennas 

and antenna arrays but care must be taken to obtain a sufficiently refined model and a full 

convergence study would be required. With HFSS such an effort may not be justified for 

obtaining better accuracy compared to a 2.5-D model such as EMP which is reasonable accurate, 

fast, and has been successfully used for the design of aperture-coupled patch antenna arrays. 

 13
 

 14

7. References 

1 EMAG Technologies Inc., EMPiCASSO, http://www.emagware.com/empicasso.html, 2006. 

2. Ansoft Corporation, High Frequency Structure Simulator (HFSS), 

http://www.ansoft.com/products/hf/hfss/, 2006.

3. Rogers Corp., Advance Circuit Materials Division Data Sheet, www.rogerscorporation.com, 

2006.

4. Weiss, S.; Leshchyshyn, A. Anechoic Chamber Upgrade; ARAED-TR-95009; Army 

Armament RDEC, May 1995. 
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