NTIA Report 96-331 

Radio Propagation Considerations for 

Local Multipoint Distribution Systems 

R.A. Dalke 

G.A. Hufford 

R.L. Ketchum 

U.S. DEPARTMENT OF COMMERCE 

Mickey Kantor, Secretary 

Larry Irving, Assistant Secretary 

for Communications and Information 

August 1996 

iii 

CONTENTS 

Page 

1. INTRODUCTION .....................................................................................................................1 

2. RADIO PROPAGATION..........................................................................................................2

 2.1 Clear Air Absorption............................................................................................................2 

 2.2 Effects of Rain .....................................................................................................................2 

 2.3 Diffraction............................................................................................................................6 

 2.4 Effects of Vegetation ...........................................................................................................7 

 2.5 Multipath..............................................................................................................................8 

 2.6 Passive Repeaters.................................................................................................................9 

3. EXAMPLES OF MEASURED DATA ...................................................................................10 

4. SYSTEM REQUIREMENTS..................................................................................................15 

5. LINK BUDGETS.....................................................................................................................17 

6. CONCLUDING REMARKS...................................................................................................21 

 

7. REFERENCES ........................................................................................................................24 

Radio Propagation Considerations 

for Local Multipoint Distribution Systems 

Roger Dalke, George Hufford, and Ronald Ketchum∗

A local multipoint distribution system will essentially broadcast television 

signals (and perhaps more) to subscribers in small cells. It has been proposed 

to put such systems in the frequency band from 27.5 to 29.5 GHz where the 

wave length is only about 1 cm, and where equipment is not well established 

and propagation effects are not entirely known. In this report we discuss what 

is known about the expected behavior of the radio waves and we suggest areas 

that need more study. 

Key words: atmospheric absorption; LMDS; millimeter waves; power budgets; rain attenuation; 

television 

1. INTRODUCTION 

It has been suggested that the band from 27.5-29.5 GHz be used for a local multipoint 

distribution system (LMDS). The notion is that one can essentially broadcast television signals 

into small cells and that the wide bandwidths available will compensate for the hazards that arise 

because of the high frequencies. Other services could also be accomodated and it has even been 

suggested that a type of two-way communication service could be established. 

Since such a system would be competing with standard broadcast stations, with cable 

installations, and perhaps also with direct broadcast satellites, its signal must be reliable and of 

high quality. An important consideration is therefore whether this is technically feasible. One 

very suspect part of this question is the over-the-air transmission channel itself, and it is this part 

that is reviewed in this report. 

In what is probably the most important scenario, the 2-GHz band would be divided into two subbands each 1 GHz wide. These in turn would be divided into 50 video channels each 20 MHz 

wide. These latter channels would carry the separate television signals. Using the present 

technology, a standard NTSC signal at baseband would be “FM’ed” into the proper subchannel. 

Assuming the video signal has frequencies up to 4 MHz, this would allow a modulation index β

of less than 1.5. 

In this same scenario, the service area of a single transmitter would be a small cell, perhaps 6 km 

in radius. If a larger area is involved, it would be divided into these smaller cells

 

∗

 The authors are with the Institute for Telecommunication Sciences, National Telecommunications and Information 

Administration, U.S. Department of Commerce, Boulder, Colorado 80303. This report was prepared as part of a 

Cooperative Research and Development Agreement between ITS and Telesis Technologies Laboratory, Inc. 
2

which are served by an alternation of the two different sub-bands and the two orthogonal 

(horizontal and vertical) polarizations. Highly directive antennas at the receivers would play an 

important role. 

2. RADIO PROPAGATION 

The commercial use of these proposed frequencies for over-the-air transmission would be 

something new, and whether the goals of the present proposal are technically achievable is not 

certain. The wave length is approximately 1 cm so that high gain antennas may be built that 

have small dimensions. But also such small wavelengths imply that almost any object in the 

radio path will have a strong effect on the transmission. In free space, the basic transmission 

loss (the loss in power between antenna terminals, assuming no-gain, isotropic, antennas) is 

Lbf = 92.45 + 20log FGHz Dkm dB

where FGHz is the frequency in gigahertz and Dkm the distance in kilometers. At 28 GHz and 6 

km this becomes 137.0 dB. This (plus the other effects we discuss below) is the loss that must 

be countered by transmitter power, antenna gains, and modulation gains. 

2.1 Clear Air Absorption 

At frequencies above 10 GHz, radio waves propagating through the atmosphere are subject to 

molecular absorption. Although frequencies near 28 GHz are in a “window”—comfortably 

between the water vapor absorption line at 22 GHz and the band of oxygen lines near 60 GHz—

there will nevertheless be some residual effects from the tails of these and other lines. Such 

effects can be evaluated using the Millimeterwave Propagation Model of Liebe [1, 2]; in Table 1 

we have displayed the resulting specific attenuation as a function of temperature and humidity. 

Note that on a hot, muggy day, a 6-km path could suffer perhaps 5 dB additional attenuation. 

Table 1. Clear Air Absorption at 28 GHz 

Relative Humidity 

Temp (C) 0% 50% 100% 

 0° 0.02 0.05 0.08 (dB/km) 

10° .02 .08 .14 

20° .02 .12 .25 

30° .02 .20 .44 

40° .01 .33 .79 

2.2 Effects of Rain 

Absorption and scattering of radiowave energy, due to the presence of raindrops, can severely 

degrade the reliability and performance of communication links. Attenuation resulting from 

propagation through rain drops is often the most significant threat to EHF
3

telecommunication availability. For many paths, predictions based on rain attenuation alone 

would be sufficient, with the error due to exclusion of the other propagation effects being much 

less than the normal year-to-year variation in rain attenuation. Note, however, dispersion 

(frequency selectivity) because of rain is not considered significant for bandwidths of less than 1 

GHz [3]. 

Rain attenuation is a function of drop shape, drop sizes, rain rate, and attenuation cross section. 

Classical rain attenuation models assume that the wave decays exponentially as it propagates 

through the volume of rain, the drops are spherical, and the contributions of each drop are 

additive and independent of the other drops. Typically, the specific attenuation through rain is 

approximated by 

α = aRb

 dB/km 

where the rain rate R is measured in mm/hr and the parameters a and b depend on the distribution 

of drop sizes and on the radio wave frequency. Tables of these parameters for several raindrop 

size distributions have been computed by Olsen et al. [4]. This approximation has been shown to 

be in excellent agreement with Mie scattering calculations for spherical drops over a wide range 

of frequencies (up to 200 GHz at 0°C for the Law and Parsons distribution). 

Several investigators have studied the distribution of rain drop sizes as a function of rain rate and 

type of storm activity. The most commonly used distributions are those of Law and Parsons 

(L-P), Marshall and Palmer (M-P), and Joss and Waldvogel (J-W). Law and Parsons, indeed, 

propose two distributions, the L-PL distribution for widespread rain (with rates less than 25 

mm/hr), and the L-PH distribution for convective rain with higher rates. In general the L-P 

distributions seem to be favored for design purposes because they have been widely tested and 

compared to measurements (see, e.g., [5]). The L-PL distribution gives approximately the same 

specific attenuation as the J- W thunderstorm distribution and the specific attenuation of the M-P 

and L-PH are approximately the same. 

Allen [6] points out that at EHF there is a range of more than a factor of 2 in specific attenuation 

for different drop size distributions used by Olsen et al. and a range of a factor of 4 for the 

different climate regions used by Dutton et al. [7]. The resulting uncertainty is the most critical 

limitation to reliable prediction of EHF system availability. This would indicate that drop-size 

distributions dependent upon climate or type of rain will need to be developed to improve 

predictions of the cumulative distributions of EHF attenuation. For the geographical regions 

under consideration in this report (San Francisco and Los Angeles, California) the low rain rate 

distributions are applicable. In some sense, the L-PL distribution provides an upper bound on 

attenuation based on commonly used drop-size distributions for low rain rates (< 25 mm/hr) and 

was used for the link budget calculations provided in this report. 

System availability (cumulative distribution of attenuation) is typically determined by using 

the Rice-Holmberg point rain rate distribution [8] and the point to path algorithm provided 

by Crane [9]. The Rice-Holmberg model gives the rain rate distribution in terms
4

of commonly recorded climatological parameters (total precipitation and the fraction of rainfall 

from convective storms). The Crane model is widely used and has been shown by Dutton [10] to 

be one of the better models. 

Data recorded at meteorological stations in Los Angeles and San Francisco were used to 

calculate the rain rate distribution and attenuation shown in Table 2 and in Figures 1 and 2. The 

attenuation is plotted as a function of path length for rain rates exceeded less than 0.1% of a year. 

Assuming a 6-km path in the San Francisco area, the results of this analysis indicate that a 

margin of approximately 15 dB is required to achieve an annual availability of 99.9% (13 dB for 

Los Angeles). 

Table 2. Cumulative Distribution of Point Rain Rates 

Based on the Rice-Holmberg Model 

Percent Rain Rates (mm/hr) 

of year San Francisco Los Angeles 

0.001 27.6 26.2 

0.01 18.6 17.3 

0.1 9.7 8.4 

0.2 7.0 5.7 

0.5 3.5 2.4 

The parameters a and b for specific attenuation provided by Olsen et al. [4] apply to spherical 

drops for which attenuation is independent of wave polarization. Coefficients for oblate 

spheroidal drops are given by CCIR [11]. At 30 GHz the horizontal and vertical polarization 

coefficients are: 

ah = 0.187, av = 0.167, bh = 1.021, bv = 1.0. 

Assuming a rain rate of 10 mm/hr, the specific attenuation for horizontally polarized waves is 0.3 

dB/km more than that for vertically polarized waves. 

Rain induced depolarization is produced from a differential attenuation and phase shift caused by 

nonspherical raindrops. The classical model for a falling raindrop is an oblate spheroid with its 

major axis canted to the horizontal and with major and minor axes related to the radius of a 

sphere of equal volume. For practical applications a semi-empirical relationship between rain 

attenuation and depolarization is provided by CCIR [12]: 

XPD = 20 log E║/E┴

 = 30 log FGHz – 10 log (0.516 – 0.484 cos 4τ) – 40 log (cos θ) – 23 log A

where XPD is the “cross-polarization discrimination” showing the relation between the 

copolarized received field E║ and the cross-polarized field E┴, and where τ is the tilt angle
5

Figure 1. Rain attenuation vs. path length for 99.9% availability in San Francisco. 

Curves represent Law and Parsons (L-PL and L-PH), Marshall and Palmer 

(M-P), Joss Thunderstorm (J-T), and Joss Drizzle (J-D). 

Figure 2. Rain attenuation vs. path length for 99.9% availability in Los Angeles. 

Curves represent Law and Parsons (L-PL and L-PH), Marshall and Palmer 

(M-P), Joss Thunderstorm (J-T), and Joss Drizzle (J-D). 
6

of the polarization with respect to horizontal, θ is the elevation angle of the path, and A is the 

rain attenuation in decibels. Figure 3 shows XPD as a function of attenuation for θ = 0 and for 

either horizontal polarization (τ = 0) or vertical polarization (τ = π/2). 

Fog results from the condensation of atmospheric water vapor into water droplets that remain 

suspended in air. There are two main types of fog. Advection fog is coastal fog that forms when 

warm, moist air moves over colder water. Liquid water content of advection fog does not 

normally exceed 0.4 g/m3

. Radiation fog forms inland at night, usually in valleys and low 

marshes, and along rivers. Radiation fog can have a liquid content up to 1 g/m3

. 

Specific attenuation for fog was calculated using a model developed at the Institute for 

Telecommunication Sciences (ITS) by Liebe [2]. Worst case calculations assuming 1 g/m3

 water 

result in a specific attenuation of 0.5 dB/km. For a homogeneous fog path of 6 km, the resulting 

attenuation is 3 dB. 

Figure 3. Rain depolarization at 28 GHz vs. attenuation. 

2.3. Diffraction 

Service at EHF frequencies will be almost entirely a line-of-sight service. Any large object that 

obstructs the path—even the house across the street—will introduce further large losses. 

Diffraction around such objects is nearly nonexistent, since the higher the frequency, the sharper 

the shadow. 

To make an actual analysis, the standard approach is to suppose that the diffracting edge can 

be represented as a perfect “knife edge.” Then there exist good approximations to
7

the theoretical solutions, and these have been validated experimentally many times. For 

example, consider a scenario in which a wave is transmitted for some distance and would reach a 

particular receiving antenna, except that it is interupted by the rooftop of a nearby building. 

Knowing the radio frequency, the horizontal distance from the diffracting edge to the receiving 

antenna, and the vertical depth below the grazing ray, it is possible to derive the consequent 

additional attenuation that the wave will suffer. In Figure 4 we have plotted such attenuations 

for a variety of plausible configurations. As shown, attenuations of 20-30 dB are obtained with 

depths of only a meter or two. 

Figure 4. Diffraction attenuation at 28 GHz as a function of the depth of a receiver below the 

grazing ray. As in the inset, d is the distance behind the diffracting edge and z is the 

depth. 

2.4. Effects of Vegetation 

A millimeter-wave link near ground level may run through vegetation over part of its path. 

Woods and forests are structurally very complex with constituents, such as leaves and pine 

needles, that are often much larger than a wavelength. In addition, the type and density of 

vegetation is likely to vary significantly over the transmission path. As a consequence it is 

expected that propagation through vegetation will cause attenuation, depolarization, and beam 

broadening. 

For several years, ITS has conducted a variety of experiments in an effort to characterize the 

propagation of millimeter-wave signals through vegetation. These experiments include the 

measurement of received strength of narrowband signals propagated through both coniferous 

[13] and deciduous vegetation [14] at 28.8 GHz. In support of this work, Schwering et al. 

[15] presented a theory of millimeter-wave propagation in woods and forests. More
8

recent work includes wideband propagation measurements at 30.3 GHz through a pecan orchard 

in Texas [16]. 

The experiments cited above were conducted in a regularly planted, well groomed stand of trees 

of the same species and about the same growth. This allowed for well-defined reproducible 

experimental conditions. Basically, it was found that there is a high attenuation rate at short 

distances into woods and a much reduced attenuation rate at large distances. In the transition 

region, substantial beam broadening occurs. For trees without leaves, the transition occurs after 

about eight trees as opposed to three trees when leaves are present. Attenuation in the foliated 

state was found to be as high as 12-20 dB per tree for the first three trees but is only about 0.5-

0.7 dB per tree at larger vegetation depths. 

Since many trees only partially obstruct a given path, Violette et al. [14] have estimated signal 

loss versus foliage depth for coniferous and deciduous trees. The foliage depth is calculated by 

tracing the direct path between the transmitter and receiver terminals on an orchard layout 

diagram and then summing the portions of the path that are intercepted by trees. Foliage depth 

can be expressed as an equivalent number of whole trees through division by the average tree 

width. 

In general, the results of these studies indicate that for the first 30 m of foliage depth, the 

increase in vegetation loss is nearly linear at a rate of 1.3-2.0 dB/m, depending on frequency; 

beyond 30 m, the curve decreases at a rate that averages only 0.05 dB loss per meter. 

Propagation through the deciduous orchard (in the foliated state) resulted in less loss than 

propagation through the conifer orchard for any given combination of frequency, transmitter 

height, and foliage depth. 

Measurements of multipath delays and depolarization are also given in the works cited above. In 

summary, impulse response measurements at 30.3 GHz showed the presence of multipath signals 

at delays as long as 15 ns for foliated deciduous trees. Measurements of depolarization in the 

conifer orchard show that at 28.8 GHz, cross-polarization discrimination is 12 dB for a foliage 

depth of 20 m and decreases to 9 dB after about 60 m. 

2.5. Multipath 

Because the wavelength is so small, the waves will reflect or scatter from almost any exposed 

object. Thus most receiver locations will almost surely be subject to multipath. The different 

multipath components, however, will arrive from different directions and a high gain antenna 

will render most of them invisible. The only components that would be seen are those that lie 

almost directly along the path to the transmitter. They might include, for example, a simple 

ground reflection. In such a case the path length difference (the difference between the path 

length of the multipath ray and that of the straight line direct ray) would be fairly small. Using a 

simple scenario for ground reflection on a 5-km path we have computed differences of perhaps 

16 cm. This amounts to a delay spread of about 0.5 ns, which is much too short to cause any 

effect such as intersymbol interference. On the other hand, this path length difference is about 

16 wavelengths; if the components are large enough, one can expect “flat” fading in which an 

entire 20-MHz channel disappears. 
9

2.6. Passive Repeaters 

Passive repeaters have been successfully used in microwave applications to overcome obstructions 

and reduce the number of active repeaters required. They also allow for more convenient 

placement of active repeaters (e.g., near roads and utilities). The obvious advantages of passive 

repeaters are that they require no access or power lines and are virtually maintenance free. 

Typically, passive repeaters are in the far field of the transmitter. When it is assumed that the 

phase and amplitude of the incident field are uniform over the repeater surface, the directivity is 

equivalent to an aperture with a uniform field. The maximum directivity (in the direction of the 

reflected wave) is given by: 

2

4 cos

λ

π A α D =

where λ is the wavelength, A is the area, and α is the angle between the incident ray and the 

normal to the repeater. The normalized power pattern is 

2

sin ( ) ⎟

⎠

⎞ ⎜

⎝

⎛ = u

u

p u

where 

θ

λ

π sin a

u =

and where a is the “effective width” equal to the actual width of the repeater multiplied by cos α. 

Here, θ is the observation angle measured with respect to the reflected ray. At 28 GHz, 

beamwidths corresponding to half and 10 dB down power points are given by 

a a

0.84 , 0.5 2θ 0 = degrees 

with the effective width a measured in meters. For a square repeater (with reflection in the 

principal plane), the area and half power beamwidth are related by: 

α

θ

cos

0.5 2 0

A =

The gain (20 log D) is given by 

G = 102 + 20log Acosα dB

which for beamwidths of more than a few degrees and incident angles of less than 40° can be 

reasonably approximated as 

G ≈ 90 − 40log 2θ 0 dB
10

relative to an isotropic radiator located at the center of the reflecting face fed with power equal to 

that accepted by the passive receiver. Thus the gain can be expressed approximately as a 

function of the desired beamwidth. 

For a given beamwidth, the required area and orientation (e.g., tilt) of a rectangular repeater are 

easily calculated. For example, if a beamwidth of 3° was desired, the reflector would need to be 

about 60 cm square, and the resulting gain would be 70 dB. 

Narrow beamwidths improve discrimination and reduce the possibility of interference. A 

potential problem associated with narrow beam repeaters is that proper alignment is difficult. 

Also, the narrow beam repeater may not provide sufficient coverage for all households in a 

shadow zone. 

3. EXAMPLES OF MEASURED DATA 

There have been many radio propagation measurements made at frequencies near 28 GHz. Most 

of them have explored situations resembling satellite-earth paths or land-mobile links in which 

path elevation angles are high or the subscriber terminal has a broad beam antenna. Some, 

however, do treat situations approximating that of the proposed LMDS. 

A report by Violette et al. [17] summarizes a series of such measurements that were designed to 

study propagation characteristics for paths between mobile terminals in urban and suburban 

environments. In particular, much of the experimental work involves narrow beam receiving 

antennas (typically 1.2°) and EHF frequencies in the range of interest for LMDS. Results from 

this study that are relevant to LMDS are discussed below. Unless otherwise indicated, the 

measurements described in this section are for narrowband 28.8-GHz signals using a 

10°-beamwidth transmitting antenna and a 1.2°-beamwidth receiving antenna. 

Reflections from various environmental surfaces are important since they may act as a primary 

signal source (nonline-of-sight) and/or an interfering source (multipath). Normal incidence 

reflection measurements involving structures composed of common building materials (e.g., 

brick or concrete) show that losses of 7-15 dB (relative to free space) can be expected. In 

addition there is significant variation depending on path length and spatial position. Presumably 

this is due to surface roughness. 

Reflections from street surfaces are another important source of reflected energy. Measurements 

of signal amplitude as a function of transmitter to receiver offset for a rural and an urban setting 

are shown in Figure 5. Here both the transmitter and receiver are near ground level. Multipath 

interference as discussed in Section 2.5 of this report is clearly evident. The rural measurements 

show deep fades between 0.2 and 0.6 km. Fading is not evident for distances of less than 0.2 km 

due to the narrow beamwidth of the receiver. The urban path also shows deep fades, although 

the pattern is quite different. This difference is most likely due to multipath interference from 

buildings or other scattering objects (e.g., traffic blocking the ground reflections). 

The strength of reflections from building walls in an urban setting were investigated by pointing 

the 10°-beamwidth transmitting antenna away from the direct path in a series
11

Figure 5. Signal amplitude as a function of range for two environments in and around Denver, 

Colorado: the top graph shows measurements along 17th Street in downtown Denver, 

the bottom along a rural asphalt road. The antennas were about 2 m above ground. 

These plots are reproduced from [17]. 
12

of 10° steps. A receiver scan (azimuthal) was made to detect reflected signals for each step. 

This more closely simulates the effects that may be expected for an omnidirectional transmitter 

and narrow beam receiver. The results clearly show significant reflected signals for both vertical 

and horizontal polarization (see for example, Figure 6). Reflections with amplitudes 10-20 dB 

below the direct ray are present. The reflections are oblique, however the amplitudes seem 

consistent with normal incidence reflections discussed above. It should be noted that the 

measurements do not include elevation scans. 

Multipath, by introducing frequency selective fading, can have a significant impact on broadband 

signals. The urban measurements [17] included multipath measurements with a 1-GHz 

bandwidth signal at 30.3 GHz. Transmitter and receiver beamwidths were 30° and 2.5°, 

respectively. Figures 7 and 8 are examples of results for receiver on-line pointing, and 3°-

receiver off-line pointing. Multipath signals with delays exceeding 10 ns are clearly evident. As 

may be expected, the off-line pointing results in a relative amplitude increase for the multipath 

components. 

Coherence bandwidth (which describes when frequency selective fading becomes important) is 

sometimes defined as the 3-dB down point of the autocorrelation function. Using the model 

proposed by Hufford [18], the coherence bandwidth is 

2πτ

1 ∆f =

where τ is the delay time. By this definition, a 15-ns delay yields a coherence bandwidth of 

roughly 10 MHz; thus, frequency selective fading is an issue for the proposed LMDS. 

Results of measurements for narrow band 28.8-GHz radio wave propagation in residential areas 

are also given by Violette et al. [17]. The environments were classified as “dense” and “sparse” 

residential, consisting of single family one-story dwellings with mature vegetation. No 

particular trend based on the neighborhood classification is apparent. Path lengths varied from 

100-1450 m. In most cases, the paths were blocked by obstacles (houses and/or trees). Resulting 

losses due to obstructions for most cases ranged from 33-60 dB, averaging at about 50 dB. 

They determined that propagation by diffraction from roof edges and especially treetops 

accounted for most of the received signal. This was indicated by a test where the loss was 

reduced about 20 dB by elevating the antenna 10°, roughly toward the treetop level. In this case 

the treetops appeared in the common volume of the antenna beams of each terminal. A similar 

situation existed in another test where large fluctuations in signal levels corresponded with tree 

motion due to wind. 

The measurements described above provide useful information regarding potential problems 

associated with the implementation of LMDS. For example, the results clearly indicate 

that multipath fading is a potential problem. The extent of the problem depends on many 

factors including transmitter and receiver locations (height is of particular importance) 

and path length and environment (suburban, urban, and rural). Most of the work cited
13

Figure 6. Measured signal levels near street level in downtown Denver, Colorado, with the 

transmitter antenna pointing away from the direct ray at angles of 0°, 10°, 20°, and 

30°. In the top graph the antennas were vertically polarized, in the bottom 

horizontally. These plots are reproduced from [17]. 
14

Figure 7. Impulse response records near street level in downtown Denver, Colorado. 

The vertical scale is 5 dB/div; the horizontal scale 1 ns/div. The graph is 

reproduced from [17]. 

Figure 8. Impulse response records near street level in downtown Denver, Colorado, at 

roughly the same location as in Figure 7. The receiving antenna is pointed 3° 

off-path. The vertical scale is 5 dB/div; the horizontal scale 1 ns/div. The 

graph is reproduced from [17]. 
15

above involved transmitters and receivers that are close to the ground and have short paths (less 

than 1.5 km). Recently, ITS conducted experimental measurements that more closely resemble 

the transmitter/receiver geometry associated with LMDS for a suburban environment [19]. A 

brief summary of the pertinent results are given below. 

Data at forty-five suburban receiver sites was collected and evaluated. Transmitters (with 26° 

beamwidth) were located at 16 and 40 m above grade with a median T-R path length of 4.5 km. 

Receiving antennas (5.5° beamwidth) were located at a height of 1 m above the roof level of 

nearby dwellings. The transmitted signal was broadband (1 GHz) at 30.3 GHz. Measured 

excess path losses varied from –6 dB to +32 dB. The median values for the 16-m transmitter 

height was 18 dB and 15 dB for the 40-m height. Multipath interference was found to be 

minimal. Measured delay spreads were between 0.7 and 10 ns with a median value of 1 ns. 

4. SYSTEM REQUIREMENTS 

For the proposed system, the received signal will need to be of high quality and high reliability, 

and therefore some amount of spectrum spreading (to trade wider bandwidths for higher signalto-noise ratios) seems necessary. While digital television with the consequent availability of 

error-correcting codes seems highly desirable, it is an advanced technique that will probably 

have to await future development. For the present, available analogue techniques still allow us 

to use wide band frequency modulation to obtain useful improvements. 

The so-called “FM improvement” is the ratio between the output SNR of the FM receiver, and 

the output SNR of an AM receiver in the same noise environment. It has the value (see [20], for 

example) 

2 ρ FM = 3β

where β is the “modulation index.” Note that the required bandwidth for the FM transmission is, 

by Carson’s rule, B = 2fm(β+1) where fm is the one-sided bandwidth of the baseband modulating 

signal. For the NTSC video signal, fm = 4.2 MHz. Note, too, that the improvement is not 

available unless the RF input “carrier-to-noise ratio” (CNR) exceeds an FM “threshold” of about 

7 or 9 dB. 

In standard audio FM transmissions, further improvements are possible by pre-emphasizing the 

higher frequencies of the signal at the transmitter and de-emphasizing them at the receiver. This 

works because most of the signal is concentrated at the lower frequencies, while most of the 

noise uses the de-emphasized higher frequencies. In the case of NTSC television, however, there 

are portions of the signal (particularly the chrominance signals) that use the upper frequencies, 

and the usual pre-emphasis/de-emphasis technique is no longer effective. We should note, 

however, that the microwave relay service and the satellite relay service both use FM modulation 

in their operations and they have developed their own pre-emphasis/de-emphasis procedure. 

This is mostly done to make television and FDM telephony appear to make similar demands 

upon the system. In the case of television, the lower frequencies are depressed relative to the 

higher frequencies. The result is a very slight improvement in SNR of about 2.9 dB (see [21], 

for example). 
16

The important criterion for a system design is to ensure that the signal-to-noise ratio is amply 

large. In the present case this means that the final picture as viewed on the screen is of adequate 

quality. This, of course, is a subjective matter; the problem has been presented to “viewing 

panels” sponsored by the 1950’s TASO group [22] that have produced results describing, for 

example, how “quality” varies with SNR. These TASO results are now widely used for design 

purposes and may be represented by the formula 

G = 7.7 – 0.16 x SNR 

where G is the picture grade as seen by the “average” viewer. Qualitatively, this picture 

grade has values 1 = excellent, 2 = fine, 3 = passable, 4 = marginal, 5 = inferior, and 

6 = unusable. In the context of the LMDS service it is proposed to require a grade 2 picture 

with a consequent required signal-to-noise ratio of 35.6 dB. Note that current television 

broadcasting systems are designed (see [23], for example) around a grade 3 picture, thus 

requiring only 29.4-dB SNR. 

As it turns out, noise in the video signal becomes, to the human eye, less important as the 

frequency increases. Thus there is a difference between the noise power as measured by a 

wattmeter and as measured by the human eye. To allow for this difference the CCIR [24] has 

recommended a “unified weighting network” to be inserted in the probe of the wattmeter used to 

measure the thermal noise. The consequence is that the noise power to be used in estimating a 

signal-to-noise ratio is considerably smaller than that due to a flat thermal noise. For an NTSC 

signal that has been frequency modulated and then de-emphasized as described above, this 

subjective noise power is about 9.9 dB below the unaltered noise (see [21], for example). 

It should be noted that at 28 GHz there is no appreciable external noise—either natural or 

manmade. The sky temperature horizontally will equal the ambient temperature so that the 

resulting received noise will be the same as simple thermal noise. The only real possibility for 

additional external noise might be from the sun. If the receiving antenna points directly at the 

sun (something that might very well happen at sunrise or sunset) it will observe an increase of 

approximately 13 dB. Otherwise, the only additional noise will arise internally from the receiver 

itself. For this we shall assume here a receiver noise figure of 6 dB, although much quieter 

receivers do exist. 

Part of the present proposal for LMDS involves a network of transmitters each serving a circular 

area of perhaps 5 or 10 km. The arrangement is thus reminiscent of cellular telephones, and just 

as in the cellular environment it is now important to consider interference between neighboring 

cells. 

The plan is to divide the 2-GHz band between 27.5 and 29.5 GHz into two 1-GHz blocks 

and furthermore to use variously, vertical and horizontal polarization from separate transmitters. 

Thus, in some sense there are four distinct “orthogonal” circuits that can be used to 

separate the cells. The two polarizations, however, provide a rather inexact orthogonalization, 

and it is problematical whether the concept would actually work sufficiently well.
17

There are a large number of places where a cross-polarized wave can be generated: reflections 

and scattering from walls, roofs, and trees—even the antennas are usually imperfectly polarized. 

The addition of oblate rain drops is only one more impairment. 

On the other hand, there is an argument by which such a cellular network might perform 

adequately: that the narrow beam antennas proposed will already provide the necessary 

discrimination. They should pick up only the transmitter they point to and ignore all others. If 

this is a valid notion one might even ask why the system needs two channels or two 

polarizations. If we assume many identically modulated transmitters using approximately the 

same frequencies and the same polarizations, then simply turning to the best available signal 

(normally the nearest) might be an effective strategy. Of course, those who live directly on the 

line connecting two of these transmitters might still have trouble. For example, if someone is at 

the edge of a cell and sees a second transmitter directly behind the assigned transmitter, then that 

second will be about three times further away than the first. On a clear, dry day, the second 

signal will be only about 9.5 dB below the first, and that is just about at the “FM capture” 

threshold. In such a case it might be useful to have even a slight addition to the desired-toundesired signal ratio as could be obtained, for example, by the use of orthogonal polarizations. 

5. LINK BUDGETS 

Of primary interest in this link budget analysis is the effect of the over-the-air transmission 

channel. As discussed in Section 2, free space loss and rain are perhaps the most important 

factors for line-of-sight (LOS) paths, both of which are a function of path length. The ratio of 

the power available at the receiver-to-reference noise power is plotted as a function of path 

length (cell radius) in Figures 9 and 10. In addition to free space loss and rain/water vapor 

attenuation, these curves include the following factors: 

• transmitter power per channel (0.0 dBW); 

• transmitting antenna gain (10 dBi); 

• receiver antenna gain (7.5′′ dish: 32 dBi); and 

• noise power (NTSC) kT0B, B = 8.4 MHz (–134.7 dBW). 

In this list, receiving and transmitting antenna gains are representative of presently available 

equipment. The noise power is referenced to 290 K for the NTSC bandwidth of 8.4 MHz. If 

there are additional factors or if there should be changes to the factors above, one can simply add 

to or subtract from the CNR for a specified path length. 

The link budget curves show the estimated CNR for a cell radius of up to 15 km using the given 

transmitter power, antenna gain, and rain or water vapor attenuation (99.9% availability) in San 

Francisco and Los Angeles. The CNR required at the receiver for FM is a function of the 

modulation index (β), receiver noise figure, and minimum SNR that will provide the desired 

picture quality. The receiver noise figure is effectively fixed by the available technology. For 

the purposes of this analysis, the SNR is set at 36 dB based on the TASO specifications for a 

“fine” picture. The modulation index can be adjusted by changing the FM bandwidth. The 

minimum CNR required for 18- and 36-MHz bandwidths are plotted in Figures 9 and 10 as 

horizontal lines (thresholds) using the following factors: 
18

Figure 9. San Francisco: Link budget as a function of path length. 

Figure 10. Los Angeles: Link budget as a function of path length. 
19

• receiver noise figure (6 dB); 

• SNR as specified by TASO (36 dB = “fine” picture quality); 

• FM advantage 3β2, 

 β = 1.14 at 18 MHz, β = 3.28 at 36 MHz; and 

• emphasis and unified noise weighting 12.8 dB. 

Other important considerations include the effects of propagation through vegetation, obstructions, and multipath fading. These effects are not included in the link budget curves shown. 

As indicated in previous sections, attenuation per tree can be as high as 20 dB. Diffraction at 

even small depths (see Figure 4) easily results in more than 10 dB of attenuation. Based on the 

measurements described in Section 3 it is expected that attenuation due to vegetation and/or 

obstructions (e.g., diffraction) will be likely at many locations in a suburban environment. Here 

one may loosely predict 20-30 dB additional attenuation for partially obstructed paths. Also, 

measurements indicate that if reflections from buildings are used as the primary path, additional 

reflection losses will likely exceed 20 dB. 

The link budget for a passive repeater can be included using formulas from Section 2.6. For 

example, assuming a 0.36-m2

 repeater, the gain is roughly 70 dB. If the repeater is 3 km from 

the transmitter, and the receiver is 0.25 km from the repeater, the CNR at the receiver is 

CNR at repeater (Figure 8 for San Francisco) 45 dB 

 (clear air 50% humidity) 

Repeater gain 70 dB 

Free Space Loss (repeater to receiver) 109 dB 

 CNR at receiver 6 dB 

A sample link budget for a cell radius of 5.6 km in San Francisco using a 1 W per channel 

transmitter is calculated as follows: 

Transmitter power/channel Pt = 0.0 dBW 

Transmitter antenna gain Gt = 10. dBi 

Receiver antenna gain, 7.5′′ dish Gr = 32. dBi 

Free space loss Lbf = 136.2 dB 

Rain attenuation Ar = 12. dB 

Water vapor attenuation at 100% humidity Aw = 1.4 dB 

Clear air attenuation at 50% humidity A0 = .7 dB 

Noise power (NTSC) kT0B, B = 8.4 MHz Nr = –134.7 dBW 

Receiver Noise Figure Nf = 6. dB 

FM improvement, 3β2

 at 18 MHz MFM = 5.9 dB 

De-emphasis and the unified noise weighting E = 12.8 dB 

Thus, for the rain-faded (99.9% availability) 18-MHz FM bandwidth, 

SNR = Pt + Gt + Gr – Lbf – Ar – Aw – Nr – Nf + MFM + E = 39.8 dB 
20

and with a clear atmosphere, 

SNR = Pt + Gt + Gr – Lbf – A0 – Nr – Nf + MFM + E = 52.5 dB 

We should recall that the required SNR (for a grade 2 picture) is 36 dB. 

Another way to look at these results is to assume that this required SNR corresponds to a 

“normal” situation. Then the proposed design can be said to provide a fade margin of 

52.5 – 36.0 = 16.5 dB, so that adequate service is provided as long as fades from rain or trees or 

other obstructions do not exceed this value. In Figure 11 we have plotted cumulative 

distributions of basic transmission loss assuming extra absorption due only to rainfall. Note that 

16.5 dB below the normal level at 6 km, service should be available except for approximately 

0.04% of the time. 

Although this analysis seems adequate for rainfall fading, there are field measurements that 

indicate there will likely be a significant number of homes in the suburban environment that 

suffer another 20 dB or more of attenuation due to vegetation or obstructions (see Section 3). If 

the above fade margin is then not sufficient, received SNR will need to be increased using one of 

a variety of techniques. One might increase the transmitter power or the modulation index. For 

example, if the bandwidth is increased to 36 MHz, MFM increases to 15 dB. 

As proposed, LMDS will be designed for two-way communications. In this case the link 

budget factors (Lbf, Ar, Aw, A0, and antenna gains) for the over-the-air channel will 

Figure 11. The cumulative distribution for the basic transmission loss on line-of-sight paths of 

various lengths. Rain statistics used are those for San Francisco. 
21

remain the same. For customer to base transmissions, it is expected that transmitter power will 

be much less and the signal will be narrow band (perhaps digital); in this case the SNR 

improvements as described above are not applicable. Other factors (such as losses due to 

obstructions or reflections) that affect the primary signal must again be taken into account. 

Additional information regarding the system (transmitter power, type of modulation, receiver 

specifications) are required to determine the feasibility of two-way communications. 

6. CONCLUDING REMARKS 

The successful implementation of LMDS depends to a large extent on factors affecting the overthe-air channel. Several of these factors, including rain, clear air attenuation, vegetation, 

obstructions, and multipath, have been reviewed in this report. A summary of the results and 

recommended measurements needed to address uncertainties are discussed below. 

Losses due to rain and humidity are reasonably well understood. Empirical and semi-empirical 

models for rain have effectively been used in satellite and ground based communications design. 

Cell size is dictated primarily by system parameters (e.g., transmitter power, modulation, and 

antenna gain) and the desired availability based on rain attenuation models. Predicting the 

number of homes that can be served based on rain attenuation is a straight-forward process. 

Figures 1, 2, 9, and 10 of this report can be used to estimate system requirements for a desired 

cell coverage area in San Francisco or Los Angeles. 

Uncertainties in predicting rain propagation losses are primarily due to difficulties in predicting 

drop size and shape distribution and the fact that rain is usually not homogeneous over a given 

path. In this report we have used the commonly accepted drop size distribution for low rain rates 

and the commonly accepted model for point-to-path rain attenuation. Note, however, that the 

latter is a function of prevailing meteorological conditions, and the resolution of uncertainties in 

the Crane global model [9] for a given region will require a series of measurements during rainy 

weather. 

Propagation losses resulting from vegetation and other obstructions are much more difficult to 

quantify and will likely vary depending on the “man made” geography. In addition, simply 

increasing transmitter power or altering cell size may not solve the problem. Experimental work 

cited in this report shows that such losses for a suburban area result in a further 20-30 dB excess 

path loss for a significant number of homes. In this investigation it appeared that attenuation was 

most likely caused by vegetation and possibly by diffraction around building edges. These paths 

were said to be “obstructed LOS” since the type of obstruction was not obvious to the 

investigators. The results are highly specific since they represent a particular type of geography 

with respect to buildings, vegetation, and terrain. At the present time these results can only be 

used as a very rough guide to estimate the severity of losses that may be encountered. 

Another important issue is the role of reflected signals in the implementation of LMDS. Two 

important questions are: “Can reflections from environmental surfaces be used as a
22

primary signal when an LOS path is not available?” and “Is interference due to reflections a 

problem for LOS paths?” As indicated in Section 3, measurements have been made that show in 

an urban environment reflected signals are clearly detectable (both at 28.8 GHz cw and 30.3 

GHz broadband). Propagation losses due to reflections ranged from 10 to more than 30 dB. In 

this case, the transmitter and receiver were near ground level and the paths were short (less than 

2 km). An important remaining question is whether or not sufficient reflected signal strength 

would be available in “shadowed” areas (with respect to LMDS transmitters) to provide adequate 

coverage for residents. 

As proposed, LMDS would use orthogonal polarization for isolation purposes. As shown in 

Section 2, rain-induced depolarization is not a major problem. Measurements performed by ITS, 

however, show that cross-polarization levels are significant (less than 10 dB down) for clear air 

non-LOS paths involving microwaves. Also, the experimental results described in Section 3 

show high cross-polarization levels when vegetation is in the path. These results indicate that 

polarization isolation may not be effective in general. 

In summary, obstructions (such as vegetation and man-made structures) provide the most 

difficult challenge in predicting the number of homes that a base station can serve. Such 

predictions are also compounded by other issues such as the amount of effort a customer will (or 

is allowed to) go to in order to receive the LMDS signal. For example, problems with vegetation 

may be eliminated if customers can (and will) place antennas above tree levels. It might be 

possible to place a common receiving antenna on the roof of an apartment complex so that 

potential customers in a shadow zone can receive the signal. Assuming that such measures will 

not be undertaken, the impact of vegetation, shadowing, and multipath on the number of 

residences that can be reached is quite uncertain at this point and will require further 

investigation. 

Future studies of the viability of proposed LMDS should include a well thought out measurement 

campaign. Ideally, the goal would be to gather enough data to create realistic statistical models 

for the prediction of received signal levels in a variety of urban, suburban, and rural 

environments. The development of such models requires one to confront difficult issues such as 

“What constitutes an archetypal urban or suburban environment?” or “How should the data be 

acquired and how much is enough?”. 

Clearly, a massive effort would be required to achieve all of the desired goal. But a more modest 

study can be started, first by identifying obvious areas where radio propagation problems are 

expected. In addition to providing information regarding the influence of these environments on 

LMDS, they will provide input needed to develop more general models. Future measurement 

efforts should include the environments and general considerations described below. 

The proposed LMDS system relies heavily on polarization to prevent interference. ITS has 

found that cross-polarization amplitudes can be significant for non-LOS paths. Further 

measurements are required to determine if the use of polarization diversity will be effective. 

Cross-polarization measurements should be performed in all of the environments described 

below. 
23

1. Urban (multistory buildings) shadow zones: 

The presence of multistory buildings in urban areas is expected to create shadow zones. In such 

cases, a large number of customers (e.g., adjacent residential neighborhoods and high-rise 

apartments) may not have LOS access to the transmitters. Potential problem areas in large cities 

that might support LMDS should be identified. Measurements (which might involve azimuthal 

scanning) to determine available signal levels should be performed. Several different locations 

should be used to obtain statistics on signal levels available in such urban shadow zones. 

2. Urban multipath fading: 

Recent measurements performed by ITS show that multipath interference was not a severe 

problem in a suburban environment consisting mainly of single story residences (see the last 

paragraph of Section 3). These results are not likely to be applicable to an urban environment 

with a high density of large structures. It is recommended that broadband measurements 

simulating LMDS transmitter/receiver geometry be performed in an urban environment and 

adjacent residential areas in order to determine the extent of multipath fading problems. 

3. Suburban (single-story dwellings) environments with mature vegetation: 

ITS has found that signal levels are strongly affected by vegetation. It is expected that the 

presence of tall mature trees will have a significant impact on signal levels for a number of 

homes. Neighborhoods with relatively high, moderate, and low densities of mature trees should 

be identified. Measurements should be made to determine estimates of the spatial distribution of 

signal levels in each case. 

4. Inclement weather: 

Measurements of signal levels for the maximum expected LOS path lengths during rain storms 

should be made to provide verification of the Crane model for the Los Angeles and San 

Francisco regions. 
24

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25

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